hi
-^
^ J ^^-^u.
^
^
^
r-
^
^
^
"oo^
'
^
^
y
y
—
0.125
■^^
/^
r"
/
/\
—
—
—
0.25
/t
f
0.5
/
"
f
2.5
0 0.5 1.0 1.5 2.0 2.5 3.0 3-5 4.0
OJ
UJQ
Fig. 1 — Characteristics of a simple variable structure.
There still remains the possibility, however, of obtaining a network
in which the distortion can be kept within tolerable limits over a given
range. The quantities Y^, Fo, and Z, which are, of course, all functions
of frequency, allow us to determine the transfer admittance at three
values of R. The transfer admittances at other settings will then be
fixed. If we suppose for simplicity, that the extreme characteristics,
corresponding to Y^ and Fo, are set by the engineering requirements on
the structure, the problem reduces to that of so choosing Z in relation
to these quantities that the distortion is as small as possible at inter-
mediate settings of R.
Of the variety of possibilities open in the selection of Z, one in par-
ticular commends itself by the simplicity and symmetry of the results
to which it leads. It is given by the condition
232 BELL SYSTEM TECHNICAL JOURNAL
Z=^^R,, (3)
where i?n is an arbitrary constant which represents, physically, a
reference value for the variable resistance. With the help of this con-
dition (2) becomes
-^(1 + n z
Y^ VFoF. ^^^^ , (4)
which can be rewritten in a slightly different notation as
where e~*, e"^", x and e'f stand respectively for the quantities F, V Fn F^,
R . Z
Ao Ao
The significance of the assumption made in equation (3) is apparent
from an inspection of equation (5). When R — Ro the total loss 6 of
the circuit is equal to ^o- The quantity do can therefore be described
as the average or reference loss of the circuit, corresponding to the
average or reference value of R. It is represented by the middle curve
shown on Fig. 2. Setting R = 0 or R — oo gives the symmetrically
located extreme curves do zt aRp
^2(a2-,)
aRp J,
2(a2-,)|
aRp
e* = a
ep+*
-J 1 1 1 Lji 1 1 I I I < I
0.5 1.0 1.5 2.0 2.5 3.0 3.5
Wp
Fig. 3 — The simplest type of symmetrical variable equalizer.
Fig. 4 — Adjustment of the variable characteristic by the addition
of an auxiliarj' network.
236 BELL SYSTEM TECHNICAL JOURNAL
The efifect of the added network is easily understood from the pre-
ceding equations. It will be noticed that although these equations
were written under the assumption that i? is a real quantity, they will
still be valid if R is complex. We need therefore merely to replace R by
the impedance of the auxiliary network terminated by the variable
resistance. If we represent this impedance by Zr, the appropriate
expression is
^^-^"l+xtanhV'' ^^^
where i^ is the transfer constant of the added network and x is, as
before, the ratio of the variable resistance to Rq. Since reciprocal val-
ues of X still correspond to reciprocal values of ^- , all of the preceding
Ro
conditions of symmetry in the resulting family of characteristics are
maintained. The simplest formulation for the new 9? is secured from
equation (6). Upon replacing the ".r" of this expression by -^- we
Rii
readily find that the equation becomes
X — 1
0 = 0'
—
e
6
4
^
--^
— -*—
__3jU— —
^^^
— —
1
^^fe
___^,____^
-<^
0.53
r-
p--<^
^-^IT"
-—o ,
_0;27
2
^^
---^
^SIT^
-
^~"^^
^Cl '^
-
0
) 200 300 400 500 600 700 800 900
FREQUENCY IN KILOCYCLES PER SECOND
Fig. 6 — Characteristics of the structure shown by Fig. 5.
238
BELL SYSTEM TECHNICAL JOURNAL
satisfy the condition expressed by equation (3). For example, any
structure having the general configuration of Fig. 7 will meet this con-
dition provided the impedances Zi, Z^ and Z3 are so related that when
the network is considered as a 4-terminal structure transmitting from
a-a' to b-h' it has a constant resistance image impedance equal to
i?o at a-a' .
In contrast to the network of Fig. 3 in which both ^0 and
^
^^
^
^
/
eo
y"
/
X
^
^'
^
^
So-*
^
^
Fig. 8 — Variable equalizers with varying reference characteristics.
A third simple network is shown on Fig. 9. Its properties are the
converse of those secured from the structures of Fig. 8. The reference
loss, ^0, is now a constant, while ^ varies with frequency in a manner
'See, for example, "Distortion Correction in Electrical Circuits," O. J. Zobel,
Bell Sys. Tech. Jour., July, 1928.
240
BELL SYSTEM TECHNICAL JOURNAL
which depends upon the choice of Z. An additional control of 0 can
of course be obtained by the introduction of an auxiliary structure in
front of the variable resistance. As in the previous example, the
illustrative curves are drawn on the assumption that the characterizing
2a^Ro
.* =
= K-57fe)
Ro^
2a
2a2-2a+i
^ — '
-—
"
©0+*
A
y^
/
r
a =0.9
/
\l
A
eo
^ '
V
\
\.
eo-t
1
0.5
1.0
2.5
3.0
1.5 2.0
OJ
LOq
Fig. 9 — A variable equalizer requiring only one general impedance branch.
impedance Z is a simple inductance. It will be observed that the
curves "see-saw," the attenuation at certain frequencies increasing
while that at other frequencies is decreased. This phenomenon
depends upon the choice of the parameter a. It disappears when a is
assigned either extreme value i or 1, and becomes most pronounced
at the intermediate value a = 1/V2. A similar effect can also be
VARIABLE EQUALIZERS
241
produced in the networks we have already considered since, as equation
(8) shows, the variable attenuation will change sign if the phase shift
of the auxiliary network is allowed to increase beyond 45°.
In the fourth structure, shown by Fig. 10, both ^o and ip are variable.
4^^^ pT
1 y^ 1
W^ z„
Ro ' '
^21 2
Z2I
1 0
ZmZoj-Ro
/
/
f
/
/
^
-^
y
y
/
y
y
eo+<^
/
A
^y^
2Ro
a'Zsi
2^21
6-^ = 1 + ^
m
:iRo
Ro
pQo
=[-^r
^11 ^21 ~ Ro
26
/
"
/
/
2" = i^Ro
/
/
/
^
/
/
/
(a
y
—
"
/
''
'^
y
^
eo-*
//
V
/eo
/;
/
/
/
/
/
7
/
00+*
/
y.
/
^^
^
y
1.5 2.0
UJ
Wo
3.5
Fig. 11 — A variable equalizer with zero loss at one frequency for all settings
of the controlling element.
elaborate and difficult to deal with. It possesses, however, the salient
property that when Zn = 0, perfect transmission of power from one
terminating impedance to the other is secured at any setting of the
variable resistance. This property suggests that the network may be
VARIABLE EQUALIZERS
243
useful for systems where it is necessary to introduce variable equali-
zation without attenuation of the channels having the lowest signal
level.
The final structure is shown in Fig. 12. Its chief point of interest
I — O— '
2Z2
iZ|
xz,
Z2
X
iz.
n
2Z2
y. 14
/
/
/
/
Z| = LujLo
Z2= ^u;oLo
/
/
©0+*
/
/
^
"
/
^
^
^
/
/
eo
^
/
^
/
^
-^
^
^
^
*\
\
"^
^^
— ■
eo-*
1.5 2.0
Wo
2.5
3.0
3.5
Fig. 12 — A variable equalizer adapted to a general controlling impedance.
is the fact that the controlling element, instead of being a variable
resistance, is a variable general impedance, which has been labelled
xZ\ in the drawing. In practical cases, of course, Z\ will ordinarily
be a simple resistance, inductance, or capacity. In addition to the
variable branch the network also includes two fixed branches propor-
244 BELL SYSTEM TECHNICAL JOURNAL
tional to Zi and three fixed branches proportional to a second general
impedance Z2. The change from a variable resistance to a variable
impedance makes little difference in the analysis. It is merely neces-
sary to replace each R and i?o by a Z or Zo in every equation. So
long as equation (3), with the appropriate modification, is satisfied,
as it is in this structure, the resulting family of attenuation character-
istics will have the same general symmetrical form as those obtained
from the resistance controlled devices. A set of curves illustrating
this point is shown at the bottom of Fig. 12. They are drawn on the
assumption that the Zi and Z2 impedances are respectively inductances
and resistances.
The structure of Fig. 12 will still function satisfactorily as a variable
equalizer if it is turned inside out so that the Zi/2 impedances become
the terminations and the central shunt branch becomes the variable.
In this event the present variable impedance must be set at its nominal
value. The resulting structure is essentially the inverse of the network
of Fig. 12. In the same way, of course, each of the other configurations
which have been described can be replaced by its inverse.
An Explanation of the Common Battery Anti-sidetone
Subscriber Set
By C. O. GIBBON
THE telephone transmitter serves to convert sound waves into
their electrical facsimile; but in performing this primary function
the transmitter also acts as an amplifier. Under some conditions the
electrical power output of a transmitter may be more than a thousand
times as great as the acoustic power activating it. Part of this greatly
augmented power is dissipated in the circuit of the telephone set; part
is impressed upon the telephone line, whence it is propagated on to
the distant listener; and part finds its way into the receiver of the
same set, where it is reconverted into sound waves. Speech or noise,
picked up by the transmitter and reproduced by the receiver of the
same set, is called sidetone.
Noise picked up and amplified by the transmitter and heard as
sidetone tends to obscure incoming speech, thereby impairing re-
ception. Similarly, the sound of his own voice, heard more loudly
than normal as sidetone because of transmitter amplification, impels
the talker involuntarily to lower his voice; thus impairing the reception
of his speech at the far end of the connection. The consequent
desirability of reducing sidetone has long been recognized, and operator
and subscriber sets which accomplish this have been developed.
Circuit schematics of the common battery sidetone and anti-sidetone
subscriber sets at present standard in the Bell System are shown in
Fig. 1. The anti-sidetone set has become increasingly common during
the past few years, and because of the improvements in effective
transmission which it affords, bids fair ultimately to be well nigh uni-
versally employed. It is, therefore, not surprising that numerous
requests have arisen for an explanation of this anti-sidetone circuit which
may be more easily followed than one based on the methods usually
employed in network analysis. The present paper provides an ex-
planation by means of diagrams with a minimum of mathematical
treatment which it is believed those to whom the mathematical approach
does not appeal will find helpful in picturing the behavior of this circuit.
The explanation given in this paper is, however, confined to idealized
conditions. No concern is given to whether the conditions necessary
to exact attainment of the balances described are actually feasible;
245
246
BELL SYSTEM TECHNICAL JOURNAL
nor is any attempt made to discuss questions of practical design be-
yond pointing out something of the nature of the problems involved.
Equations for this anti-sidetone circuit are given and discussed in an
appendix, and a vector diagram is shown which illustrates graphically
relations among the currents and voltages under the ideal condition of
exact balances.
Rearrangement of Circuit Patterns
Simplified explanations of anti-sidetone sets are most frequently
based upon analogues with balanced arrangements resembling the
SIDETONE CIRCUIT (S)
ANTI-SIDETONE CIRCUIT (a)
\
WINDING
:^ WINDING
\
-N \ NET- \
^ B
WINDING
d\ \W0RK ^
^\ N ^
>
> A
/
>LINE
\^(V
6y; ^
> L
yx/
X/RECEIVER
R
TRANSMITTER'-r^M
t
i-\>j
Fig. 1
I-ig. 2
3a -i{^^ = I2a
I OJM^
lA
I
\ B
-+•
'••3A
^(1)
^3A
XS «M] '^
Fig. 9
Group III — Components of transmitting E.M.F.'s and assumed mesh currents
250 BELL SYSTEM TECHNICAL JOURNAL
when transmitting (see Fig. 5 of Group III), it will be looked upon as
the same passive impedance in tandem with an impedanceless generator
whose e.m.f. equals the variations in voltage drop (of the battery sup-
ply current) across the transmitter due to the changes in its impedance
which occur when the transmitter is agitated by sound. This e.m.f.
thus replaces in the circuit the sound engendered variations in the
transmitter impedance, thereby permitting this impedance to be
treated as a constant. Being impedanceless, this generator may,
without effect upon the circuit, be replaced by two impedanceless
generators — each having the same e.m.f. as the first — connected in
parallel as shown in Fig. 6. The direct connection between points
a and h, however, is shunted by the impedanceless path ach, so that the
direct connection ah may, without effect, be broken as in Fig. 7.
Hence, the two equal e.m.f. 's in Fig. 7, acting simultaneously, are
equivalent to the single e.m.f. in Fig. 5; and the mesh currents in the
two figures are, therefore, identical. Hereafter, Fig. 7, rather than
Fig. 5, will be considered the transmitting condition.
This transmitting condition may, however, be broken into two com-
ponent conditions. By the fundamental principle known as the Super-
position Theorem, the currents in Fig. 7 are equal to the sum of the
currents which would result from each of the two e.m.f.'s acting alone.
In other words, the transmitting currents in Fig. 7 are equal to the sum
of the corresponding currents in Figs. 8 and 9. But by a second
fundamental principle called the Reciprocity Theorem, the current at
any point Z in a circuit, due to an e.m.f. at any other point Y, is equal
to the current which would result at Y from an equal e.m.f. at X.
Applying this to Figs. 8 and 9, in which Ei = £2, the mesh currents
pointed out by the arrows joining these two schematics are equal,
viz.:
Ifl = I^s and m = m. (1)
Of the above components of the transmitting currents, those in
Fig. 9 are due to an e.m.f. acting in mesh 1, i.e., in series with the line
impedance. This, however, is also the condition when receiving, as
will be seen by comparing Figs. 9 and 4.
Neutralizing Balance — ^Receiving Efficiency
Consider next the purpose of winding C. It is, of course, desirable
that the transmitting and receiving efficiencies be undiminished by the
anti-sidetone arrangement. If it is possible so to adjust the couplings
among windings A, B and C that the current If^ in Fig. 4 or 9 is
zero, the balancing branch can then be disconnected without effect
COMMON BATTERY ANTI-SIDETONE SUBSCRIBER SET 251
E| - E2
T(l2)lr(l)+TC2)('BY THE SUPER-
'1"; ■ IS IS ]^ POSITION THEOREM
U,^,
BY THE RECIPROCITY
THEOREM AND £[=£2
,(!)_' (1)1 DUETO^r(l)_,
'•IA-|MS|_WINDINGCj ^2A ^
r(l2)i,(l) + r(2) / BYTHESUPER-
I lA — 1|A^^IA \POSITION THEOREM
FIGURE
8S
FIGURE
9S
IGURE
9A
>!y FIGURE
E| = E2
Group IV — Transmitting efificiency — sidetone balance
252 BELL SYSTEM TECHNICAL JOURNAL
upon the currents, and the circuit will thereby be reduced to the side-
tone circuit. Hence, with this ideal adjustment of the couplings, the
receiving efficiency of the anti-sidetone circuit will be the same as
that of its sidetone complement.
Although equally satisfactory designs could be worked out with
other polings of the coil windings, the relative inductive directions
among windings A, B and C in the circuit here dealt with are such
that, if a current were passed through all three in series, windings
A and B would be inductively aiding; and C would be inductively
opposed to both A and B. Returning to the above condition for
maintaining the receiving efficiency, namely, that I'i\ be made zero,
the windings of the coil must be so adjusted that the sum of the
two voltages induced in winding C by its inductive couplings with
windings A and B is equal and opposite the voltage drop across the
receiver. With the windings poled as just stated, this requires that
(+ Pp^Z^^c) + (- mZBc) = -{- I'ilZn). (2)
This voltage balance expressed by eq. (2) will be referred to as neutral-
izing balance: its attainment requires the coil windings be adjusted to
meet the relation shown by eq. (6) in the appendix.
It is important to note that neutralizing balance, and hence the
efficiency relations which depend upon it, are independent of the
impressed e.m.f. £i, of the line impedance Zl and the self-impedance
of winding A , and of the network impedance Z.v and the self-impedance
of winding C. This of course follows from the fact that none of these
quantities is involved in eq. (2).
Transmitting Efficiency "-
It will now be shown that the transmitting efficiency of the anti-
sidetone circuit is the same as that of its sidetone complement; and
that this equality, like that of the receiving efficiencies, results from
the neutralizing balance effected by winding C. This is true if, with
equal transmitter e.m.f.'s in the top and bottom diagrams of Group IV,
the line currents are equal, viz., if
7(12) _ Ti\2)
To prove this relation, refer to Fig. 7 A at the bottom of Group IV and
move up step by step to Fig. 9A, observing the relations between
mesh currents indicated by the arrows. It will be seen that
COMMON BATTERY ANTI-SIDETONE SUBSCRIBER SET 253
But in Figs. 9A and 95, due to neutralizing balance, as was shown in
discussing receiving efficiencies,
r(i) _ n\) 3n^ ni) _ r(i)
Finally, continuing from Fig. 95 upward to Fig. 75, it is seen that
J- 1.S I -'2S ~ -'IS •
Hence,
ni2) _ ni) I /-(i) _ ni) _i_ rd) _ 7'(i2)
-^lA — -'lA T^ -'2.4 — -'is T^ -'25 — -* IS •
Primary Purposes of Winding C and of Neutralizing Balance
The above relations between the efficiencies of the anti-sidetone
circuit and those of its sidetone complement are, however, merely
incidental to the primary purposes of winding C and of the neutralizing
balance which it provides. The major purpose of winding C is that,
entirely apart from its neutralizing action, the voltages induced in it
through its couplings make it possible to obtain sidetone balance by
adjusting Zn] i.e. — referring to Fig. 1A at the bottom of Group IV^ —
given any value of Zl, it is theoretically possible so to adjust Zn that
P^^ = — I^^^^K The current through the receiver under the trans-
mitting condition, i.e., sidetone, will then be zero. Neutralizing
balance permits this adjustment of Zn to be made without affecting
the circuit efficiencies.
Sidetone Balance
The following discussion of sidetone balance will proceed on the
assumption that the couplings of winding C with windings A and B
have already been adjusted for neutralizing balance, since this con-
dition is required to maintain the circuit efficiencies. This approach
is merely a matter of convenience, however, for it will be indicated that
the impedance of TV required to effect sidetone balance is the same
whether the neutralizing balance is taken into account or ignored.
Although sidetone balance is made possible by the couplings of winding
C, and the impedance of N needed to reduce sidetone to zero does de-
pend upon the values to which the self and mutual impedances of this
winding have been adjusted, neither the attainment of sidetone balance
nor the value of Z^ required to provide it depends upon the existence
of neutralizing balance.
If sidetone is to be zero, the voltage across the receiver under the
transmitting condition, i.e., the sum of the voltages across C and N,
must be made zero. Expressed in terms of the voltages in Fig. 7 A
at the bottom of Group IV, this requires that
n^rzAc - i'h'Zbc - nTiZc + z^) = o. (3)
254 BELL SYSTEM TECHNICAL JOURNAL
Here, at once, the dependence of sidetone balance upon the presence
of the inductively coupled third winding is apparent. Without
winding C the impedances Zac, Zbc and Zc in the above expression
would all be zero, the terms in which they occur would drop out, and
the requirement for elimination of sidetone would reduce to Zn = 0,
i.e., a short circuit across the receiver.
The remainder of this discussion of sidetone balance can be carried
out more conveniently in terms of the mesh currents than in terms of
the above voltages. As already noted, the voltage balance just
examined is equivalent to requiring that I2A and I'^f '^^ Fig. lA be
made equal and opposite. But I2A is the sum of the two components,
I2A ii^ Fig- 9^ and I2A in Fig. 8^ ; and, because of neutralizing balance,
-^3A^ = If A in Fig. 8^1. Furthermore, by the Reciprocity Theorem,
the component I2A always equals /j^^ ; and the latter, like the former, is
independent of Zn- The condition for sidetone balance may, there-
fore, be expressed in terms of the currents in Fig. 8^ as
Ifl + ITa + IfA = 0. (4)
The question, then, is whether N can be so adjusted that the sum of
the three mesh currents in Fig. 8^ is made zero; and it is fairly evident
such an adjustment for any given value of Zl is theoretically possible.
Since /j^l is known to be independent of Zn, and because inspection
shows the circuit to be symmetrical with respect to L and N, it appears
that I^A must be independent of L — an intuitive inference which the
Reciprocity Theorem confirms. The value of I2A depends, of course,
upon both Zl and Zn. Hence, with the value of /j^l remaining fixed
as Zn is varied, and with I'iX independent of Zl but under the direct
control of Zn, it may be concluded possible to meet eq. (4) by a suitable
choice of Zn for any given value of Zl- The value of Zn required to
attain sidetone balance is shown by eq. (7) in the appendix.
With N so adjusted that sidetone is zero, it is obvious the receiver
impedance may be changed in any way whatever without upsetting
the sidetone balance. The same is true of any change in the im-
pedance of the transmitter; because this, being equivalent to a com-
pensating change in the transmitter e.m.f., would cause all mesh
currents to change in the same proportion ; thus leaving the balance
expressed by eq. (4) undisturbed. Hence, the impedance of N re-
quired to provide sidetone balance is independent of the receiver and
of the transmitter. But as has already been seen, the couplings of
winding C necessary to provide the neutralizing balance in eq. (2) do
depend upon the receiver and transmitter impedances. The sig-
nificance of this observation is that although the value of Zn required
COMMON BATTERY ANTI-SIDETONE SUBSCRIBER SET 255
to provide sidetone balance does depend upon the values of the coup-
lings of C, neither the attainment of the balance in eq. (4) nor the
impedance of N required to provide it depends upon the balance in
eq. (2) being met. In other words, the neutralizing balance expressed
by eq. (2), and the sidetone balance expressed by eq. (4), are mutually
independent; either may be attained without the other.
Practical Considerations
With the simple types of coil and network permitted by economic
and space limitations, the balances upon which the above performance
of the anti-sidetone circuit depends can be obtained exactly only with
a given line and at a single frequency. For practical purposes, how-
ever, exact balances are needless. Sound leakage under the receiver
cap and conduction through the head structure fix a limit beyond which
further reduction in sidetone is not of value. Actual designs, therefore,
aim at the best compromise in reducing sidetone over the voice range
and the range of line impedances important in practice, as judged by
the resulting effective transmission performance obtained with the
instruments employed. Under typical plant conditions, designs now
in service reduce the volume of sidetone with present instruments to a
level averaging around 10 to 12 db below that of the complementary
sidetone sets.
APPENDIX
Algebraic Solution of Circuit Equations
Referring to Fig. 10, the following circuit equations may be written:
-7* -
-AC-
.. Zg ^-— Zbc—-
+-
Fig. 10 — ■* The poling of windings A and B is series aiding. Winding C is
poled in series opposition to windings A and B.
{Z L-\- Z A-\- Z t) I iA-\- {Zt — Zab)J2a— ZacIza = Ei
{Zj'—Zab)IiA-{-{Zt-\-Zb-\-Zr-\-Zs)I2A-\- {Zii-\-ZBc)IiA =E2
— ZacIia-\- {Zr-\-Zbc)I2a-\-{Zr-\-Zc-\-Z.w)I3a = 0
(5)
256
BELL SYSTEM TECHNICAL JOURNAL
M
fe
U '^ O'O
oj-c ■; rt
>H S2 u
u 0) O) 0) ^
>>
I
COMMON BATTERY ANTI-SIDETONE SUBSCRIBER SET 257
These cover both transmitting and receiving conditions: when trans-
mitting, El = E2\ and when receiving, Ei — 0.
The relation which the induction coil must meet in order to provide
neutralizing balance can be determined by solving eqs. (5) under the
receiving condition £2 = 0, and imposing the requirement that /3'J = 0.
This gives as the relation to be met.
ZaC ZaB — Zt />.n
(6)
Zbc + Zr Zt + Zb + Zr + Zs
In like manner, the value of Zn needed to provide sidetone balance
can be determined by solving eqs. (5) under the transmitting condition
El = £2, and imposing the requirement that I2A + -^3^^ = 0. This
value of Z\, regardless of whether or not eq. (2) is imposed as a further
condition in its derivation, is found to be
r^ ry V \ Z ,\ c{Z A C 4" ZbC " ZaB " Zb " Z ^) ,_,
ZjV — ^BC — Z.C -\ ^ j y 1 7^ • \l )
^ L -r ^A -\- ^AB
Note that Zn is here independent of Zt and Zr, except as these may
enter implicitly as factors affecting the impedances at right in de-
signing the coil for optimum performance with specified instruments.
In other words, the transmitter and receiver may be changed without
disturbing the sidetone balance. Such a change would, however,
upset the neutralizing balance, thereby altering the efficiencies from
those of the sidetone circuit.
Vector Diagram
Relations among the component mesh currents in an anti-sidetone
circuit of this type under ideal conditions of exact neutralizing and
sidetone balances, are illustrated by the vector diagram in Fig. 11.
As all of the current vectors indicate current per volt impressed, those
for the mesh currents under the receiving condition in Fig. AA are
identical with those under the component of the transmitting condi-
tion in Fig. 9A. Vector sums of the mesh currents show the current
through the receiver and that fed into the line when transmitting, and
illustrate the sidetone balance. Vectors of the three voltages acting
around the third mesh in Figs. 4^ and 9 A are also shown, together
with their summation. The latter illustrates the neutralizing balance
of eq. (2).
The Occurrence and Effect of Lockout Occasioned by
Two Echo Suppressors
By ARTHUR W. HORTON, Jr.
"The Time Factor in Telephone Transmission" by O. B.
Blackwell (B. S. T. J. January 1932) deals with a number of
problems which arise in connection with telephone circuits having
long transmission times. This paper discusses one such effect,
the occurrence of lockout caused by the echo suppressors involved
in a long telephone connection.
The occurrence of lockout is shown to cause an increase in
repetition rate, which is ordinarily small for circuits as now used
commercially. The increase in repetition rate is approximately
proportional to the number of lockouts occurring and to their
mean duration, or to the per cent of time locked out.
The expected number of lockouts is shown to depend upon the
characteristic time intervals of conversational speech, the relay
hangovers, the delay of the circuit and location of the echo sup-
pressors with respect to the ends of the circuit. Subject to certain
restrictions, the expected number of lockouts increases with the
delay included between the echo suppressors, and is nearly inde-
pendent of the delays between the suppressors and the circuit
terminals.
The mean duration of lockouts is shown to be proportional to
the relay hangovers.
Introduction
WHEN carrying on a conversation over a telephone circuit of
moderate length, the subscribers are ordinarily unaware of any
limitations imposed upon the free interchange of information. As the
length of the circuit is increased the time factor ^ becomes increasingly
important and may become manifest in a number of ways. One result
of the time factor is the occurrence of echoes which become apparent
when the speech energy reflected from the end of the circuit is delayed
in returning to the talking subscriber. When the circuit is equipped
with an echo suppressor to render this efifect unnoticeable, or when a
long connection of two such circuits is made, the action of the suppres-
sors is such as to make the circuit inoperative in the opposite direction
to which speech is being transmitted. Consequently the subscribers
are no longer able to interchange information with the ease and rapidity
that would be enjoyed on a shorter circuit.
1 "The Time Factor in Telephone Transmission," O. B. Blackwell, Bell System
Technical Journal, January 1932.
258
THE OCCURRENCE AND EFFECT OF LOCKOUT 259
A circuit equipped with a single echo suppressor is always operative
in one direction, and although both subscribers may start to talk at
about the same instant, one or the other will always obtain control of
the circuit and his speech will be heard by the other subscriber. The
principal difficulties encountered on circuits of this type become ap-
parent when the hangover times of the relays are large. There is some
difficulty in interrupting since the relays do not release during the
pauses between words, and a quick response following a pause by the
first talker may reach the suppressing relay before it has released, result-
ing in a mutilation of the initial part of the response.
When two echo suppressors are used, as is the case when two circuits
each equipped with an echo suppressor are connected in tandem, similar
difficulties may be encountered. In addition, lockout, or blocking of
transmission in both directions, may occur and may persist for an ap-
preciable time. Since neither subscriber is aware that the other is
talking, both may continue talking until one or the other of the relays
releases during a pause and enables the circuit in the appropriate direc-
tion. Thus neither subscriber will be conscious of the fact that a
lockout has occurred unless he realizes from the context that some part
of the conversation has been lost.
This paper discusses the manner in which lockouts can occur, and
presents the results of a series of tests to determine their effect upon
conversation as measured by repetition rate.^ These results indicate
that the repetition rate increases with the per cent of time during which
lockout occurs. It is shown that the locked out time can be approxi-
mately calculated in terms of the circuit constants and suitable charac-
teristic intervals of conversational speech, and the calculated values
can in turn be used to predict the effects of lockout on repetition rate.
In terms of the effect upon the talkers, a lockout may be considered
to occur when speech currents from one talker are prevented from
reaching the other talker by one of the suppressing relays and those
same speech currents operate another suppressor in such a way that
speech currents from the latter talker are prevented from reaching the
former. This description of lockout should not be considered as a
precise definition since it does not specify the duration of a lockout.
No definition in terms of measurements made upon speech at the circuit
terminals would be free from difficulties in practical application, such
as that of determining with sufficient precision the instants at which
speech is considered to start and stop, and that of determining the
direction of transmission. A definition in terms of the operations of
^ "Rating the Transmission Performance of Telephone Circuits," W. H. Martin,
Bell System Technical Journal, January 1931.
260 BELL SYSTEM TECHNICAL JOURNAL
the suppressors is somewhat simpler to formulate, but may be difficult
to apply when the echo suppressors are separated geographically. In
the tests to be described there was no such separation involved and
consequently the operation of the suppressors could be readily ob-
served and easily and accurately measured. Accordingly for the pur-
pose of this paper we shall define a lockout as the condition in which the
suppressors are operated in such a way that both directions of trans-
mission are simultaneously blocked. In general, lockouts may be
caused by speech, or noise, or both, but the term will be used here to
apply to the case in which operations of the suppressors have been
caused by speech from both ends of the circuit.
In the course of a conversation the interchange of speech is ordinarily
such that the circuit is alternately disabled by the two suppressors in
one direction or the other depending upon the direction of transmission.
When a pause of sufficient duration occurs, the party not in control of
the circuit may reply at such a time that he obtains control of the echo
suppressor nearest to his end of the circuit, and a lockout can occur
provided that his speech does not reach the distant suppressor until
after the party formerly in control of the circuit has resumed talking
and has obtained control of that suppressor. The occurrence of lock-
out is therefore dependent upon the time intervals in conversational
speech and upon the constants of the circuit.
The Manner in Which Lockout can Occur
The characteristic time intervals of conversational speech upon which
the occurrence of lockout depends, are treated in a companion paper by
Mr. Norwine and Mr. Murphy.^ It is sufficient here to define two
such characteristic intervals based on a simplified concept of a conver-
sation. Neglecting grammatical considerations we can consider speech
to be composed of a sequence of vocal intervals defined and separated
by silent intervals. The lengths of these silent intervals will be called
resumption times. Likewise a conversation may be considered to be
composed of an alternate succession of speeches, defined and separated
by intervals, the lengths of which will be called response times. An
ambiguity occurs when both parties talk simultaneously but, for the
purpose of this discussion, it will be sufficient to allow for this situation
by admitting negative response times.
Figure 1 represents a generalized four-wire circuit equipped with two
echo suppressors located at different distances from the ends of the
circuit. The transmission times of the different parts of the circuit are
3 " Characteristic Time Intervals in Telephonic Conversation," A. C. Norwine
and O. J. Murphy, this issue of the Bell System Technical Journal.
THE OCCURRENCE AND EFFECT OF LOCKOUT
261
indicated on the figure with appropriate subscripts and the two direc-
tions of transmission are differentiated by the primed and unprimed
notation. The suppression points are indicated by arrows which repre-
sent an opening of the transmission path when the relays, or other sup-
/w+'''+Te=T
w —
— E
Tw+r + re=T
Fig. 1 — Schematic of generalized four-wire circuit equipped with two
echo suppressors.
pression devices are operated. The suppressing relays are specified
by a notation which refers either to the particular relay or to its hang-
over, or releasing time. According to the definition given above a
lockout exists during the time that both the relays he and hj are
operated.*
With the exception of the beginning and end of the conversation the
occurrence of lockout can be described in terms of the resumption and
response times following a pause by one talker, and the constants of
the circuit. Referring to Fig. 1, and considering the sequence of events
following a pause by E, we shall see that two types of lockout can occur.
The first type, which is the one usually met in practice, can occur
when he < hw + r, and hv> releases after he. A response by W and a
resumption by E are necessary to produce a lockout. It will persist as
long as both E and PF continue to talk and for an additional time equal
to the delay from the end of the circuit to the first relay to release after
a pause by one talker, plus the hangover time of that relay. A lockout
of this type may be termed a lasting lockout.
The second type can occur when hw -\- t < he, and A«, releases before
he. It is possible for a response by W to arrive at hu, and operate hj
before he has released thus causing a lockout wihch will be terminated
when he releases. A lockout of this type, which may be termed a
releasing lockout, can occur without a resumption by E, or if E's
resumption reaches hJ after /?« releases. If a releasing lockout has oc-
^ Also, according to the definition, when both the relays hy, and hJ are operated,
a condition of no practical importance.
262
BELL SYSTEM TECHNICAL JOURNAL
I
THE OCCURRENCE AND EFFECT OF LOCKOUT 263
curred and Ks resumption operates he before W's response can operate
hj , a second lockout which will be of the lasting type, will at once occur.
Otherwise W's response will operate hJ giving control of the circuit
to W.
Experimental Conditions and Data
To obtain experimental data of the occurrence of lockout in long
distance conversations and to determine the resulting effect on repeti-
tion rate, added delay and echo suppressors were inserted at the New
York end of a circuit to Chicago, Illinois. This circuit is used as a
tie line by the Western Electric Company for the transaction of com-
pany business between its Hawthorne plant and New York office.
The regular echo suppressor usually associated with the circuit at
Pittsburgh was removed for these tests. The circuit arrangement em-
ployed is shown schematically in Fig. 2, the added equipment being in-
cluded between the dotted lines. This equipment was adjusted to have
zero insertion loss and the frequency characteristic was equalized to
within ± 2 db from 200 to 3000 cycles. The overall net loss from toll
board to toll board was 7 db. The suppressors were 44-A echo sup-
pressors operating at a sensitivity of 31 db referred to the zero level
point of the circuit, except in those cases specifically mentioned. The
added delay circuits were of the acoustic type consisting essentially of a
suitable length of brass pipe terminated by high quality loud speaking
telephones together with the necessary amplifiers and equalizers to give
zero loss over the frequency range from 200 to 3000 cycles. These de-
lay circuits were available in units of 0.023, 0.05, 0.08, 0.10 and 0.15
second, and various combinations of these delays were used together
with the tie line delay of 0.043 second to obtain the circuit conditions
which were tested.
The details of the recording mechanism are indicated schematically
in Fig. 2. A relay was added in series with the shorting relay of each
echo suppressor, so that every operation of the echo suppressor relay
was accompanied by an operation of the added relay. The simultane-
ous operation of these relays energized two other relays, one of which
in turn operated a message register to record the number of lockouts,
and the other connected a 20-cycle oscillator to a cycle counter to record
the locked out time.
Service observers at New York monitored both directions of the
conversation and recorded repetitions and other pertinent data regard-
ing each call.
The circuit conditions tested are shown in Table I, the notation of
which corresponds to Fig. 1. For each condition the first line re-
264
BELL SYSTEM TECHNICAL JOURNAL
fers to transmission from west to east and the second line from east to
west. The hangovers are those of the relays which short the indicated
transmission path, for example, the figure 0.186 in the first line of Table
I refers to the hangover of the relay at the west end of the circuit which
shorts the transmission path from west to east. The designation in
Table I indicates the grouping of conditions for observation. All con-
ditions having the same numeral in the designation were observed con-
currently, the procedure being to observe 25 calls on condition a, then
25 calls on condition h and so on. In this way seasonal variations and
uncontrolled effects at the terminals or in the transmission line have been
minimized for a group of conditions bearing the same numerical desig-
TABLE I
Condition
T
Tw
hu,
T
he
Te
la
0.139
0.139
0.043
0.043
0.186
0.200
0.073
0.073
0.146
0.146
0.023
0.023
\h
0.193
0.193
0.043
0.043
0.186
0.200
0.100
0.100
0.200
0.200
0.050
0.050
\c
0.293
0.293
0.043
0.043
0.186
0.200
0.150
0.150
0.300
0.300
0.100
0.100
la
Same as la
Ih
0.139
0.139
0.043
0.043
0.186
0.200
0.073
0.073
0.200
0.200
0.023
0.023
2c
0.139
0.139
0.043
0.043
0.186
0.200
0.073
0.073
0.300
0.300
0.023
0.023
3a
Same as Ic
3b
0.316
0.316
0.043
0.043
0.186 0.250
0.250 0.146
0.023
0.023
4a
Same as
Ic, suppressor sensitivities 28 db
U
Same as
\c, suppressor sensitivities 31 db
Ac
Same as
\c, suppressor sensitivities 34 db
5a
Same as
\c, suppressor sensitivities 31 db
5b
Same as
Ic, suppressor sensitivities 41 db
6a
Same as
Ic, suppressor sensitivities 47 db
la
0.116
0.116
0.043
0.043
0.136
0.150
0.050
0.050
0.146
0.100
0.023
0.023
n
0.116
0.116
0.043
0.043
0.136
0.170
0.050
0.050
0.210
0.146
0.023
0.023
THE OCCURRENCE AND EFFECT OF LOCKOUT
TABLE I (Continued)
265
Condition
T
Tw
K
T
h.
Te
8a
0.116
0.116
0.043
0.043
0.136
0.150
0.050
0.050
0.146
0.146
0.023
0.023
Sb
0.116
0.116
0.043
0.043
0.136
0.170
0.050
0.050
0.210
0.096
0.023
0.023
9a
0.093
0.093
0.043
0.043
0.136
0.050
0.000
0.000
0.036
0.150
0.050
0.050
9b
0.093
0.093
0.043
0.043
0.136
0.150
0.000
0.000
0.136
0.150
0.050
0.050
9c
0.093
0.093
0.043
0.043
0.136
0.250
0.000
0.000
0.236
0.150
0.050
0.050
10a
0.116
0.116
0.043
0.043
0.136
0.100
0.050
0.050
0.100
0.096
0.023
0.023
106
0.193
0.193
0.043
0.043
0.136
0.150
0.100
0.100
0.150
0.150
0.050
0.050
lOr
0.293
0.293
0.043
0.043
0.136
0.150
0.150
0.150
0.250
0.250
0.100
0.100
11a
0.116
0.116
0.043
0.043
0.186
0.186
0.023
0.023
0.186
0.186
0.050
0.050
\\h
0.216
0.216
0.093
0.093
0.286
0.286
0.023
0.023
0.286
0.286
0.100
0.100
\\c
0.296
0.296
0.093
0.093
0.286
0.286
0.123
0.123
0.286
0.286
0.080
0.080
Notes
The values of delay in the column headed Tw include the delay of the tie line
and the added artificial delay.
Circuit ib arranged with relays he and hy, to short the echo suppressor without
shorting the transmission path.
nation. Observations were also made from time to time on the tie line
without added delay and with a single echo suppressor of special design.
Since this condition was not subject to lockout, these observations may
be used to give an indication of the seasonal effects, and to correct data
obtained from the different groups of tests. These data are designated
by the letter n.
The data recorded consist of the duration of each call, the number of
lockouts per call, the total locked out time per call, and the number of
repetitions per call. Calls having a duration less than 100 seconds are
not included in the data. Table II gives the number of calls observed,
the mean duration of the calls in seconds, the number of lockouts per
266
BELL SYSTEM TECHNICAL JOURNAL
100 seconds (L/lOO), the per cent of time locked out, or locked out time
in seconds per 100 seconds (LT/lOO), the number of repetitions per 100
seconds (i?/100), and the number of repetitions per 100 seconds cor-
TABLE II
Condition
Number
of Calls
Mean
Duration
Seconds
L
100
LT
100
R
100
R'
100
\a
275
451
2.61
1.13
0.40
0.40
16
275
437
2.63
1.40
0.44
0.44
U
275
456
3.68
2.34
0.53
0.53
\n
275
401
0.36
la
200
439
2.62
1.03
0.36
0.36
Ih
200
410
2.21
1.19
0.47
0.47
2c
200
393
3.86
1.54
0.45
0.45
ia
300
424
3.61
2.34
0.51
0.51
36
300
397
5.90
1.85
4a
275
457
3.26
1.63
0.53
0.51
46
275
413
3.34
1.76
0.55
0.53
4c
275
432
3.13
2.02
0.55
0.53
4»
75
396
0.38
5a
275
436
3.99
1.99
0.60
0.53
56
275
425
4.03
2.37
0.56
0.49
5n
250
392
0.43
6a
50
391
8.34
4.26
0.72
0.67
6n
125
390
0.41
la
200
410
3.00
0.79
0.52
0.41
lb
275
387
4.26
1.23
0.55
0.44
In
75
395
0.47
8a
150
387
2.51
0.74
0.49
0.36
86
300
401
4.64
1.30
0.53
0.40
8w
100
393
0.49
9a
150
413
0.83
0.02
0.43
0.36
96
150
403
2.59
0.17
0.42
0.35
9c
150
380
5.42
1.00
0.44
0.37
9n
175
399
0.43
10a
300
401
3.84
0.91
0.46
0.44
106
300
405
4.46
1.78
0.44
0.42
10c;
300
420
5.28
2.66
0.54
0.52
lOw
175
439
0.38
11a
300
413
1.64
0.64
0.42
0.37
116'
300
408
0.94
0.56
0.38
0.33
He
300
430
2.99
2.28
0.56
0.51
llw
300
396
0.41
rected for seasonal variations {R' ji^Q). These rates are obtained by
dividing the total number of occurrences, or locked out time by the total
duration for each test condition.
Effect of Lockout on Repetition Rate
It would be reasonable to expect that the repetition rate would de-
pend not only on the lockout rate, but also on the duration and type of
lockout. Considering the data as a whole there does not appear to be
THE OCCURRENCE AND EFFECT OF LOCKOUT
267
any definite relation between the lockout rate and repetition rate, al-
though in most cases an increase in lockout rate results in an increase
in repetition rate. If we exclude from consideration those cases in
which the lockouts are of very short duration and in which releasing
lockouts occur, the data indicate a somewhat closer dependence of repe-
tition rate on lockout rate. This suggests that the increase in repeti-
tion rate caused by lockouts may be proportional to the duration of
lockouts and to their frequency of occurrence, or to the per cent of
time which is locked out. Fig. 3, which shows the repetition rates.
0.7
10
1 0.6
O
O
lU
10
§0.5
a
UJ
Q.
,.^-^
,.-"
y
,'-'
k'
O £f^
-.-^'
^y"'
0
<'
J>-^
0 .
z "-^
o
\- <
l-
UJ
Si 0.3
0.2
^<.
o ^^
P.-^"
y
1.5 2.0 2.5 3.0 3.5
PER CENT OF TIME LOCKED OUT
Fig. 3 — Observed variation of corrected repetition rate with per cent of
time locked out.
corrected for seasonal variations, plotted against the per cent of time
locked out, indicates a reasonable agreement with this assumption.
The correction is applied by subtracting from the observed repetition
rate the difference between the observed repetition rate for the appro-
priate reference or n condition and a rate of 0.36, arbitrarily chosen as
equal to the lowest repetition rate observed on any of the n conditions.
All of the data are included in this figure. The dashed lines are drawn
to include all the data and have a slope estimated as average from
considering the data in individual groups. The variability in the data
may in part be attributed to the variation in the distribution of lockout
durations, since if two distributions have the same mean value but
different spreads, the lockouts comprising the distribution which in-
cludes a greater number of long lockouts might be expected to have a
greater effect upon the repetition rate. With due allowance for the
variability of the data. Fig. 3 indicates that the repetition rate
increases proportionally with the per cent of time locked out except
268 BELL SYSTEM TECHNICAL JOURNAL
possibly for values less than 0.6 per cent. The slopes of the boundary
lines are such as to show about 0.1 increase in repetition rate with each
1 per cent of locked out time, and this relation appears to hold for re-
leasing as well as lasting lockouts, and for lockouts which may be caused
by relay operations by noise.
Certain qualifications are necessary in considering the significance of
this result. The indicated increase in repetition rate may be partly due
to other causes than lockout, as for example the effects introduced by
the delay of the circuit, or by the relay hangover, during changes in the
direction of speech transmission which are not accompanied by lockout.
The net effect of these causes increases with circuit changes which in-
crease the per cent of time locked out. Consequently, the latter may
be taken as a criterion of the total effect, even though the contribu-
tion of the former to the repetition rate may be appreciable.
No general significance can be attached to the absolute values of the
repetition rates observed in these tests since it is well known that repe-
tition rates will differ for identical circuit conditions used with different
terminal conditions and by different classes of telephone subscribers.
These observed rates are significant only for comparing the relative
performance of circuits under the particular conditions of use pertain-
ing to these tests.
The significance of the results obtained depends upon the assumption
that a change in lockout which causes an increase in repetition rate is
an undesirable change and the transmission performance is thereby
degraded. In the case of certain circuit changes which introduce
changes in intelligibility the resulting changes in repetition rate can be
used to determine effective transmission ratings,^ expressed in db, of
the circuits under consideration. A corresponding procedure might
be applied to express the observed changes in repetition rate due to
lockout in terms of db, but in the absence of data to establish the
equivalence of the ratings for different types of degradation, it has not
seemed advisable to do so.
Locked Out Time in Terms of Circuit Constants
Since these tests indicate that the repetition rate is proportional to
the per cent of time locked out we can limit our consideration to the
latter as a suitable criterion for measuring the relative merit of circuits
equipped with two echo suppressors. To determine the per cent of
time locked out we can measure it directly, as has been done in these
tests, or it can be calculated in terms of the circuit constants by deter-
* "Scientific Research Applied to the Telephone Transmitter and Receiver,"
Edwin H. Colpitts, Bell System Technical Journal, July 1937.
THE OCCURRENCE AND EFFECT OF LOCKOUT
269
mining the average number of lockouts per hundred seconds and the
average duration of lockouts in terms of the circuit constants and ob-
taining the per cent of locked out time as the product of these two
quantities.
The average, or expected number of lockouts per hundred seconds
can be approximately determined from the circuit constants and the
distributions of response and resumption times. It is shown in the
appendix that, subject to certain assumptions, the probability of lock-
out following a pause is given by
^ = \ ] P^^^^ ^2(.v) dx dy,
(1)
in which pi(x) dx and piiy) dy are the probabilities that, following a
pause, the resumption time will be between x and x -\- dx and the re-
V) 12
r
\ RESPONSE
\ TIMES
T
/ r
s
1
1
\
\
\
I
1
V
\ RESUMPTION
\ TIMES
\
\
>
\
/
\.
\
>»_
.
y
/
■^
■»«.^
"^—
0.5 1.0
TIME IN SECONDS
Fig. 4 — Observed distribution of resumption and response times.
sponse time will be between y and y + dy. As suitable approximations
to these probabilities we may take the observed distributions of re-
sumption and response times. Mr. Norwine and Mr. Murphy, in their
accompanying paper,^ give distributions of resumption and response
times which are shown in Fig. 4. These distributions are expressed in
terms of the total number of resumptions, or responses and conse-
quently the data are an approximation to the conditional probability
that if a resumption or response has occurred, the resumption or re-
sponse time will be between / and / + dt. The use of these data will
^ Loc. cit.
270
BELL SYSTEM TECHNICAL JOURNAL
therefore result in calculated values of the probability of lockout which
are proportional to the desired probability, and if the value of the
integral calculated from their data is p, then
P = kp,
(2)
where ^ is a constant of proportionality which depends on the average
number of pauses occurring, and which can be determined by comparing
observed and calculated results.
The observed lockouts per hundred seconds plotted against the cal-
culated probability of lockout for each circuit condition are shown in
Fig. 5 for lasting lockouts and in Fig. 6 for releasing lockouts. The
.-'
P
,p
y'
^-»
^jr"
*'''
n ^
^
X
cr"'
y
8
o
-#
y
^fi
9
^- '
.,'-' ^
0 0.05 0.10 0.15 0.20 0.25 0.30 0.35 0.40 0.45 0.50
CALCULATED PROBABILITY
Fig. 5 — Observed lasting lockouts vs. calculated probability.
data are separated in this way, since the factor of proportionality be-
tween observed and calculated results is found to be different for the
two cases. This is probably due to the fact that the method of deter-
mining response and resumption times was such that some of the nega-
tive and shorter positive response times could not be detected. An
increase in the number of these response times would result in an in-
creased probability of releasing lockouts, which would tend to bring the
two sets of data into agreement. Greater accuracy might be obtained
if the distribution of response times were to be more accurately deter-
mined, but the present data are sufficient for approximate calculations.
In both figures the data obtained in a single group of tests are con-
nected by dotted lines. The solid lines are the best estimates to repre-
THE OCCURRENCE AND EFFECT OF LOCKOUT
271
sent the two complete sets of data. In the case of Fig. 5 the solid line
was determined by the method of least squares, omitting the data of
group 10. This omission appears to be justified since these data are
consistent among themselves and yield a factor of proportionality
2 3.5
."; 2.0
/
/
P,
/
//
/
^
yj^
/
P
'>^
/
^y^
/
y
y/
/
^
/
,//>
P
/
/
y
Z/
y
y
/
f
y/
y^
/
y
y
y J
y
y
y
/
/
P
y
^/z
f
/
/
y
f<'/
' if
y
/
y
/ y
/
/
/
/
y
y y
^ y'
;:
/
^ /
y
y
/
/
/
v/
y^
y
y
/ /
^ /
y
y
Xy
y
/
/^ .^
y
'^
/
y
y
y
<
V
y
If
y
y
y
/
y
y
/
0.10 0.15 0.20
CALCULATED PROBABILITY
Fig. 6 — Observed releasing lockouts vs. calculated probability.
which is consistent with the rest of the data, and since it is known that
many uncontrolled factors may influence the results of one particular
group of tests. In the case of Fig. 6 the solid line was obtained by
averaging the slopes and constant terms of the individual dotted lines.
Both sets of data indicate that about 0.9 lockout per hundred seconds
occurs when the calculated probability of occurrence is zero. This is
undoubtedly due to non-synchronous action of the suppressors, caused
by slight variations in sensitivity, changes in effective hangover caused
by changes in sensitivity and by occasional relay chatter. This con-
272
BELL SYSTEM TECHNICAL JOURNAL
elusion is confirmed by the tests made with no delay between the sup-
pressors, in which lockout is obviously impossible with synchronous
action, but in which lockouts were actually obtained in amount con-
sistent with the rest of the data.
Figures 5 and 6 show that the number of lasting and releasing lock-
outs can be calculated from the circuit constants and the distributions
of response and resumption times. Approximations which are suffi-
9 0.7
<^ 0.6
o
,^
/
L
/
\ .
^
<
K
^
i
D
o
/"
y
r
" c
v
^
y
^
y^
0.2
0.1
0 0.1 0.2 0.3, 0.4 0.5 0.6 0.7
tie + I^W IN SECONDS
Fig. 7 — Observed length of lockout as a function of relay hangovers.
cient for practical purposes for calculating the number of lasting and
releasing lockouts are respectively
Li
Lr
0.9 + 11.0 />,
0.9 + 17.7/?.
(3)
(4)
The duration of a lockout is obviously dependent upon the way in
which the subscribers talk and upon the hangovers of relays //,„ and hj .
Lockouts of several seconds duration have frequently been observed but
most frequently the duration of lockouts appears to be short and de-
termined primarily by the relay hangovers. Figure 7 shows the mean
duration of lasting lockouts plotted as a function of the sum of the relay
hangovers, he + hj . The straight line is the least square representa-
tion of the data, which is
Di = 0.002 + 1.16 {h, + hj).
(5)
The constant term in this equation can be neglected for approximate
calculations.
THE OCCURRENCE AND EFFECT OF LOCKOUT
273
0.50
T = 0.45
<
a = 0.1
O.AO
^
0.35
^
^
0.30,
^^
>>^
■
0.25y
/
/
^^
\
0.20 J
l^
■-—
^
/ ^
7
^^
^
0.10
/
^
-^
/
0.05>
L
■
■
0.05 0.10
0.20 0.25 0.30 0.35 0.40 0.45 0.50
T IN SECONDS
Fig. 8 — Calculated probability of lockout as a function of total circuit delay.
Since the duration of lasting and releasing lockouts is not the same,
releasing lockouts being of short duration, and since the two were not
separately observed, the data are insufficient to determine the duration
of releasing lockouts directly. However, tentative calculations indi-
cate that approximate results can be obtained by assuming that the
274
BELL SYSTEM TECHNICAL JOURNAL
mean duration of releasing lockouts is about one-quarter that of lasting
lockouts or a
D,. = 0.29 (//. + hj). (6)
0.6S
T=0.50y^
0.60
0.65
OAo/
/
a = 0.1
/
0.50
0.45
/
/
/
0.30 >/
/ /
^
/
0.40
0.35
/
/
y
//
^
l/
/
0.25/^
/
/
/
0.30
0.25
/
/^
:y
/
/
y
^
//
/
y
0.15^
/
y'^
^^
/
V'
y
^
0,10
//
V/
/y
y
^
^^
0.075
/ / /
/ >
^
1,—*-"
//,
^
/
^
■
0.050
M
i^^^
=^==
"^
0.025
0
0.2 0.3
0.5
T
0.8 0.9
Fig. 9 — Calculated probability of lockout as a function of the ratio of the delay
between the suppressors to the total circuit delay.
With the above relations between probability of lockout and circuit
constants, number of lockouts and probability of lockout, and mean
duration of lockouts and relay hangovers, it is possible to determine the
i
I
THE OCCURRENCE AND EFFECT OF LOCKOUT 275
per cent of time locked out from the circuit constants. Since this is
proportional to the repetition rate, a measure of relative circuit per-
formance is obtained in terms of the circuit constants.
As an example of such calculations let us assume that t,„ = tJ
= Te = T,.', T — T and T = T\ and /i,„ = h,^ = 2t,„ + a. Then the
constants of integration determined in the appendix become,
a = a,
b = a -\- T - T,
c = a + 2" -f T,
and the probability of a lasting lockout is proportional to
"a+T+T n^ r'y+T+T
P
Xa r^a+T+T n^ r'y+T+T
p2{y)dy I pi(x)dx + I p2iy)dy I pi{x)dx. (7)
Values of this probability for a = 0.1 are shown in Fig. 8 as a function
of the transmission time T with r, the delay between the suppressors as
a parameter and in Fig. 9 as a function tJT with 7" as a parameter.
The curves are not extended beyond T — 0.5 since smaller values are
thought to cover the range of practical interest. Furthermore, for
large values of T there is some evidence that the effect of the transmis-
sion time would be noticed by the subscribers with a consequent change
in the distributions of resumption and response times.
These curves indicate that for a constant value of r, the delay be-
tween the echo suppressors, there is little change in the probability of
lockout as the total delay of the circuit T is increased, and for a constant
value of T the probability of lockout is approximately proportional to t.
To continue with a more specific example, let us consider a telephone
connection consisting of two four-wire circuits each equipped with an
echo suppressor in the center of the circuit as shown in Fig. 10. In the
notation of Fig. 10 the relay hangovers are each equal to r + 0.100
and the constants of integration are
a = 0.100,
b = 0.100 + r,
c = 0.100 + 3t.
Since the two circuits are assumed equal, only lasting lockouts are
theoretically possible and the curves of Fig. 8 may then be used to
determine p in terms of r as defined by equation (7), which in turn may
be used to determine the expected number of lockouts from equation
(3). The mean duration of lockout is obtained by inserting the value
276 BELL SYSTEM TECHNICAL JOURNAL
of the relay hangover in equation (5) giving
D = 0.234 + 2.32 r.
The product of D with the expected number of lockouts per hundred
seconds is then equal to the per cent of time locked out, which is shown
in Fig. 10 as a function of t. By using the relation shown in Fig. 3 a .
3.5
9 0.55 -
SO.50
!}J0.40
-
l^ __ J
^ r -^.
T -<
>
y
/"
k
h
^
^ ^ ^
ECHO / ^ _
SUPPRESSOR f^ '
\
/
/
y
^
y
^
^
2.5Q
2.0 (u
2
0.04 0.06
0.08 0.10 0.12
T IN SECONDS
Fig. 10 — Calculated per cent of time locked out, and repetition rate for the indicated
circuit conditions.
second scale is shown in Fig. 10 to give the relation between the repeti-
tion rate and the delay between the suppressors. This curve shows
that the repetition rate increases with the delay between the suppres-
sors, at a gradually increasing rate up to a delay of about 0.09 seconds,
beyond which the impairment increases linearly with the delay.
Summary
It has been shown that two types of lockout, lasting and releasing
lockouts, may occur in telephone connections involving two echo sup-
pressors, and the manner of their occurrence has been discussed.
The results of an experimental investigation show that the occur-
rence of lockouts causes an increase in repetition rate, which is ap-
proximately proportional to the per cent of time locked out.
There has been presented a theoretical method for calculating the
expected number of lockouts in terms of the circuit constants which de-
THE OCCURRENCE AND EFFECT OF LOCKOUT 277
pends upon the characteristic time intervals in conversational speech.
The values which have been calculated with experimentally determined
constants are shown to agree with the observed values.
The average duration of lockouts has been found to be proportional
to the hangovers of the relays effective in lockout.
Since the per cent of time locked out is equal to the product of the
average number of lockouts per hundred seconds and the average dura-
tion of lockouts, it may be determined in terms of the circuit constants,
and used as one of the criteria of the relative performance of the circuits
under consideration.
Specific examples of such calculations have been used to illustrate the
relations between the expected number of lockouts and the circuit con-
stants, and between the repetition rate and the constants of a particular
circuit configuration.
Subject to certain restrictions on the relations between the circuit
constants, it appears that the number of lockouts and the resulting in-
crease in repetition rate are approximately proportional to the delay
included between the echo suppressors.
In conclusion I wish to express my appreciation to my associates who
have contributed to this study; in particular to Dr. G. R. Stibitz who
first developed the theoretical approach to the problem, to Mr. W. R.
Bennett and Mr. B. D. Holbrook who have contributed to the extension
of this approach, and to Mr. A. C. Norwine and Mr. O. J. Murphy who
obtained the distribution functions used in the calculations and con-
ducted the experimental work.
Appendix
To assist in formulating an expression for the probability of lockout
a number of simplifying assumptions have been made, as follows:
Pauses in speech are sufficiently separated to be considered as
independent events, or in other words the sequence of events oc-
curring at one pause have no effect upon those occurring at another.
Following a pause each speaker can start speaking only once and
only one of three events can occur.
1. The original speaker regains control of the circuit.
2. The other speaker obtains control of the circuit.
3. Lockout occurs.
Resumption and response times are independent.
The distributions of response and resumption times are inde-
pendent of the delay of the circuit and the disposition of the echo
suppressor.
The operate times of the suppressors are sufficiently small to be
neglected.
278 BELL SYSTEM TECHNICAL JOURNAL
Let pi{t)dt be the probability that the speaker in control of the circuit
will resume speaking in the interval t to t -{- dt after pausing and let
p2{t)dt be the probability that the speaker not in control of the circuit
will start speaking^^in the interval t to t -\- dt after hearing the other
speaker pause. In the latter case / may be negative. Then the proba-
bility that, following a pause, a resumption will occur in the interval
X to X -\- dx and a response will occur in the interval y to y -\- dy \s
given by
p\{x) pi{y) dx dy,
and the probability of lockout following a pause is given by
P = j j pi(x) piiy) dx dy,
in which the integration is to be performed over the region in the xy
plane which contains those values of x and y for which lockout occurs.
Assuming that either subscriber is equally likely to have control
of the circuit at any instant, the probability of a lockout following a
pause by either party is given by the average of the probabilities for
the two parties.
In determining the limits of integration there are three cases to be
considered. Assuming a pause by E
I. //,. < //,„ + r,
II. //,„ + r < //„ < h,, -i- T -\- t',
III. /;,„ + T + r' < h.
In case I only lasting lockouts can occur while in cases II and III
both lasting and releasing lockouts can occur. Case I will be used to
illustrate a method of determining the limits of integration which can
also be applied to cases II and III for which the results will be stated
without proof.
In Fig. 11 which is based on the circuit of Fig. 1, time is represented
horizontally and distances vertically, upward from the central line,
which represents the east end of the circuit, for transmission from W
to E and downward for transmission from E to W. Consider a
pause by E occurring at x = 0 at ^. The line ABDG represents the
transmission of this pause to W. The point H obtained by projecting
G to the top line determines the point y — 0. The points C and F
represent the instants at which he and /?„, release. If E resumes in
the interval AI, the resumption will arrive at the input of hw before
hw has released as determined by the point F, and E retains control
of the circuit. If, on the other hand W responds at any time prior to
THE OCCURRENCE AND EFFECT OF LOCKOUT
279
/ the response will be blocked by /?,,. until the time represented by K
and it will then be transmitted to the end of the circuit as shown by
the line JKLM. If now E resumes at any time after P the resumption
Fig. 11 — Time relations in four-wire circuit for determination of limits of integration.
will be blocked by hj since H^will have obtained control of the circuit.
A resumption in the interval IP will result in lockout since W will
control hiJ and E will control hr. If W responds at some time after /,
say at Q, a similar argument can be used to show that E must resume
in the interval SR to cause lockout. If we let
/// = a,
AI = b,
AP = c,
it can be shown that
AS - 3' - a + b,
AR = y — a + c,
and therefore the region of integration is defined by
b < X < c,
a -\- b < X < y
— -x < y < a,
a -\- c, a < V < '^ ,
in which a is the time interval the speaker not in control must wait
after hearing the other speaker pause in order to enable the response
to get through the circuit; b is the time interval after the speaker in
control pauses, during which he can gain control by resuming, regard-
less of what the other speaker does; c is the time interval the speaker
in control must pause in order to make it possible for the other speaker
to get a response through the circuit.
280 BELL SYSTEM TECHNICAL JOURNAL
These constants have the values
a = Kc — {tw + r,„'),
h = h^,
C = hu, + (r + r').
Case II. By the same method the regions of integration are
determined to be, for lasting lockouts
y < a, b < X < c,
y < a, y — a-{-b
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w
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i.
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K
Fig. 2 — Typical sections of oscillographic records.
a. Circuit reversals. New York completed talkspurt, heard Chicago's reply,
and began another talkspurt.
b. Reply by New York during pause by Chicago caused lockout. New York
gained control of the circuit.
c. Short reply by Chicago during pause by New York caused lockout. New York
regained control of the circuit.
d. Negative response time. Chicago replied before the completion of New York's
talkspurt.
TIME INTERVALS IN TELEPHONIC CONVERSATION 287
A few samples from the original oscillograms are shown in Fig. 2.
The speech energy in each sample is shown on traces 3 and 4 counting
from the top down, the upper being from Chicago and the lower from
New York. The cyclic waves on traces 2, 5 and 6 indicate respectively
lockout, establishment by Chicago and establishment by New York }
These waves were obtained from an oscillator which was concurrently
used to drive an escapement-type electric clock for measuring the total
call duration.
The top oscillogram was selected to show the simplest type of con-
versational interchange. It will be seen that New York had been talk-
ing but had reached the end of his talkspurt as marked on the film.
Approximately 0.4 second later Chicago responded, his talkspurt ap-
parently consisting of three syllables, whereupon after a further time of
about 0.35 second New York responded and continued talking. The
second film was selected to show a less simple type of interchange where-
in a long pause within a talkspurt prompted the listener to reply. In
this instance the times were such that a lockout resulted. Since the
remainder of the talkspurt by the original talker, Chicago, was short
and the responding party. New York, continued talking, the circuit
was established in New York's direction after the lockout. In the
third oscillographic strip Chicago attempted to interrupt, and a short
pause by New York permitted lockout to occur; Chicago did not gain
control of the circuit. This is an example of concurrent talkspurts,
both of which were included in the data. The fourth example was
chosen to illustrate a negative response time. In this case Chicago
began to reply before the end of New York's talkspurt; no lockout
occurred, but the first part of the reply was inaudible to New York due
to continued establishment of the circuit in the opposite direction.
It may be noted in Fig. 1 that speech from Chicago was recorded 0.25
second before it was heard by New York and that speech from New
York was recorded 0.193 second before it arrived at Chicago. Likewise
the beginning of each response did not occur at the time shown on the
oscillograms but at a time previous by the delay from the talker's
position to that of the recording means. To obtain the response times
as previously defined each apparent response time was given an ap-
propriate time correction.
Data Obtained
The more detailed observations were made on fifty-one calls with a
total recorded duration of a little over 13,000 seconds. At the record-
ing speed of 20 feet per minute this resulted in about 4400 feet of
* An establishment by a talker is said to occur when his speech energy has gained
control of all voice operated equipment in his transmission path.
288
BELL SYSTEM TECHNICAL JOURNAL
oscillograms. In all cases recording began at the start of the call, but
in some instances recording was stopped before the termination of the
call due to lack of recording paper in the oscillograph. The oscillo-
grams, ranging in length from 29.6 to 660.8 seconds, represented ob-
servations on calls whose mean duration was 430.5 seconds. The
speed of recording was such that the time intervals under observation
could readily be measured with a precision of db 0.005 second. The
conversational elements were measured with this precision and listed
in their order of occurrence for each call. The records for all calls were
10 CD
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1
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1
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/r
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MEDIAN
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1
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Fig. 3 — Lengths of talkspurts.
70
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20
then consolidated and retabulated in terms of the number of instances
of each element whose duration could be included within each of a
regular progression of time increments. For all three items of data
time cells 0.10 second wide were chosen. The data, when thus
cellularized, provided the basis for the construction of histograms from
which the time-distribution curves were obtained. These distribution
curves and their respective summation curves are given in Figs. 3, 4,
and 5. Some of the statistically significant quantities ® are tabulated
on the opposite page. The values are time intervals in seconds.
Since most telephonic speech syllables are shorter than 0.3 second
the modal value of 0.25 second for the length of talkspurts makes it
clear that monosyllabic replies are by far the most numerous. From
^ The moie is the value which occurs most frequently, i.e., the peak of the dis-
tribution curve.
The median is that value above and below which equal numbers of observations lie.
The mean is the arithmetic average of all the values observed.
TIME INTERVALS IN TELEPHONIC CONVERSATION
289
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TIME IN SECONDS
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298 BELL SYSTEM TECHNICAL JOURNAL
pigeonhole for hydrogen should contain three mass-values ; that for
helium, two; that for tin, no fewer than ten. It would make the table
impossibly crowded to print them all in this way, and consequently I
have broken it up into sections, of which Fig. 2 represents the first six
elements.
1 H O O O
2 He O O
3 Ll
4 Be
6 C ^ O
Fig. 2 — Isotopes of the first six elements.
In this figure each element has a row to itself, and each value of
mass has a column to itself, and each circle represents a stable kind of
atom. I now introduce the technical term "isotope" to distinguish
the different kinds of atoms common to a single element. Hydrogen,
you see, has three stable isotopes (there is some doubt about the
stability of the third, though none about its existence); helium two
(again there is doubt about the stability of one) ; lithium two, beryllium
only one, boron two, and carbon two of which the second will appear
in the next figure. The unit of mass is a very small amount, about
1.67 -10"-* of one gram. I do not pause to give it as accurately as I
might, for we are not going to be concerned with very exact mass-
values in this talk. The masses of the isotopes are not exactly integer
multiples of this unit; for instance, those of the three kinds of hydrogen
atoms are 1.008, 2.016 and 3.017. The departures from integer
multiples are, however, small, as you see in these three cases. Small
as they are, they are mightily important; but it is permissible to ignore
them for the purposes of this lecture, and I am going to ignore them
from now on. I will, however, speak of the integers at the heads of
the columns as "mass-numbers" rather than "masses."
Since the isotopes of an element differ in mass, what is it then that
they have in common? I answer this question by describing Ruther-
ford's second great achievement, the "nuclear atom-model." Ruther-
ford was the first to prove that the atom consists of a positively-
charged nucleus surrounded by a swarm of negative electrons. The
RADIOACTIVITY— ARTIFICIAL AND NATURAL 299
nucleus is much more massive than the electrons, and this is one of the
reasons for comparing the atom with the solar system, in which the
sun is much more massive than the planets which perpetually swing
in orbits around it. A less hackneyed and newer simile is that of
Bragg, just now, by the way, appointed as Rutherford's successor in
the Cavendish chair at Cambridge, who likens the atom to a man's
head with a swarm of gnats buzzing around it. Normally — that is to
say, when the atom is complete and electrically neutral — the negative
charges of all the electrons put together just balance the positive
charge of the nucleus. If Z be used to stand for the number of
electrons in the normal neutral atom, and —e for the charge of the
negative electron, then -\-Ze is the amount of the charge on the
nucleus. Z is called the "atomic number" of the element in question.
13 14 15 16 17 18 19 20 21 22 23
o
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t
1
^
o
o
t
I
o
o
t
o
1
i
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10 Ne
11 Na
Fig. 3 — Isotopes of the elements numbered 6 to 11.
This it is, of which all the isotopes of any one element have the same
value. This it is which distinguishes an element, and is common to
all of the different atoms of that element whatever their masses may
be. For hydrogen it is 1 ; for helium 2; for lithium 3; for uranium 92.
Each row in Fig. 2 and Fig. 3 is marked on the left not only with
the chemical symbol of the element to which the row belongs, but also
with the atomic number thereof.
Radioactivity is a feature of the nucleus. This accounts for some of
its remarkable aspects, which greatly surprised the world of physicists
and chemists when they were first established. Being a quality of
the nucleus, it varies from one isotope * to another of any element,
much more drastically than does the mass. Being a quality of the
nucleus, it is immune to the physical state of the atom — -i.e. it is the
* Isotopes were first distinguished from each other (by Soddy) by virtue of their
differences in radioactivity.
300 BELL SYSTEM TECHNICAL JOURNAL
same whether the atom is part of a solid, a liquid or a gas; it is immune
also to the chemical state of the atom, i.e. it is the same whether the
radioactive element is isolated or is part of a chemical compound. It
is also immune to heat and to all of the many other agencies which
physicists and chemists have at their command.
Radioactivity being a feature of the nucleus, every chemical symbol
which I use from now on will refer to the nucleus of an atom and not to
the atom as a whole. "Be" will stand for beryllium nuclei, "F" for
fluorine nuclei, "AI" for aluminium nuclei. For most elements,
though, there are two or more different sorts of nuclei distinguished
from each other by their masses, and the symbol must tell us which is
meant. The custom is to write the mass-number of the isotope in
question as if it were an exponent: H' and H'^ and H^ for the three kinds
of hydrogen nuclei, He^ and He* for the two kinds of helium nuclei,
Li® and Li'^ to distinguish between the isotopes of lithium, and so on.
If in addition one wants to remind the reader of the atomic number,
one writes it as a subscript before the chemical symbol: iH\ iH-, 2He'*,
oF^^ and the like. Purists object that either the chemical symbol or the
value of Z is superfluous when both are given, but others often like to
see them both. And now for some names: there are three nuclei which
have names of their own. The Greek words for "first" and "second"
are applied to iH^ and iH-'; they are the proton and the deuteron. The
name for 2He* is alpha- par tide; this nucleus is indeed the particle which,
as Rutherford discovered long ago, makes up one of the three kinds
of rays which radioactive bodies emits, and there never was a greater
piece of good fortune in language than that whereby this all-important
particle received the name of the first letter of the Greek alphabet, for
indeed it is the alpha of modern nuclear physics. And now another
reference to masses: the mass of the electron (when not moving ex-
tremely fast) is only about .0005 of the mass-unit which is being used
throughout this talk, and therefore the mass-numbers at the heads of
the columns in Figs. 1 and 2 and others are about as good approxima-
tions to nuclear masses as they are to atomic masses, and I shall use
them as such.
Now let us notice not only the circles of Figs. 2 and 3, but the stars
as well. The stars also stand for nuclei, but these are radioactive —
or unstable, two words which have practically the same meaning when
applied to a nucleus. At least one star appears in every row, the first
in Fig. 2 excepted. If the figure had room for ninety-two rows, one
for each element from hydrogen to uranium, there would appear at
least one star in every row below the first, excepting three (atomic
numbers 61, 85, 87) for which no isotope either stable or unstable
RADIOACTIVITY— ARTIFICIAL AND NATURAL 301
known with certainty and the element itself must still be regarded as
missing. (Furthermore there should be at least three more rows
numbered 93, 94 and 95, and containing stars but no circles.) This is
what is meant by saying that every known element, hydrogen alone
excepted, has at least one radioactive form.
Figure 2 show^s that at the beginning of the Table of the Elements,
the stable types of nuclei outnumber the unstable ones. The pre-
ponderance is gradually shifted as Z increases, and Fig. 4 exhibits to
us how greatly the radioactive nuclei outnumber the stable ones among
the elements of which the atomic numbers range from 81 to 84. In-
deed the circle which is lowest and most to the right in Fig. 4 represents
the most massive and most highly charged of all the stable nuclei
which are known (it is the solitary isotope of bismuth, atomic num-
ber 83 and mass-number 209). All the rows after 83 are occupied
entirely by stars. ^
203 TO 218
8' o o o ^ ^ -^
62
83
8A _!£_ _v_ v _V \/ \/ w
Fig. 4 — Isotopes of the elements numbered 81 to 84.
As the title of this talk has already suggested, the radioactive nuclei
are of two classes: the "natural" and the "artificial," the types already
existing in the rocks of the earth and the types made in the laboratory
by physicists employing the art of transmutation. Nearly all of the
natural types lie beyond 80 in atomic number, and most of them were
discovered in the first fifteen years after Becquerel found the first.
Two of them are identical with two of the man-made types. Apart
from these two, every one of the artificial types is a creation of the
years since 1933. One guesses that while the natural radioactive
bodies may be many, the artificial ones must surely as yet be few;
how surprising then to learn that while there are some forty of the
former, the latter after four brief years already number two hundred
and thirty! Unlike the natural ones, these artificial isotopes are
sprinkled liberally throughout the whole of the Table of the Elements,
from the second onward to the end. Not only in number but in di-
^ It must though be admitted that some of the heaviest nuclei, though demon-
strably radioactive, may exist for hundreds of millions of years before they dis-
integrate; "instability" is indeed a very relative concept!
^
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302 BELL SYSTEM TECHNICAL JOURNAL
versity of mass and charge are the artificial radioactive bodies now out-
standing by far.
These artificial examples forming now so large and important a
group among the radioactive nuclei, I make a digression to speak of
some of the transmutations from which they are derived. The art
of transmutation is already so huge a subject that the digression must
be severely limited if ever we are to come back to radioactivity. I
must therefore make only a passing allusion to the fact that the first
of the new radioactive nuclei were made by bombarding various light
elements with very energetic alpha-particles. Here the second Curie
generation must be introduced, for the daughter and son-in-law —
Irene Curie and Frederic Joliot — of the first Curie pair were the ones
who made this discovery. (It was not their entry into the field of
radioactivity, they having already studied natural radioactive bodies
for a number of years).
Returning to Figs. 2 and 3, notice that many of the radioactive
isotopes lie just one step to the right of stable isotopes: Li*, Be^", B^^,
C^*, N^^, O^^, F^", Ne^* are the examples found in these two pictures
alone. It seems as though they might differ from their neighbors
on the left — ^Li^, Be^ and so forth — by possessing an extra particle of
mass (approximately) 1 and charge zero. If only one could find such
particles roaming freely about in Nature, might one perhaps succeed
in adding them to the stable nuclei of lithium and beryllium and
boron and the other elements, and so produce these radioactive nuclei?
Such particles may indeed be found roaming about in Nature, but
not of their own volition. These "neutrons" — for such is their
name — -must themselves be set free by the art of transmutation. Free
neutrons were first produced by bombarding certain elements with
alpha- particles; the discovery was an international one, and its story
is interesting, but to keep this digression within bounds I must again
content myself with giving the names — Bothe and Becker in Germany,
Curie and Joliot in France, Chadwick in England — of those who carried
it through its consecutive stages from first intimation to triumph.
More than a hundred different ways of freeing the neutron are already
known, but of all this diversity I will take one only, which consists in
projecting deuterons against deuterons.
The "deuteron-deuteron reactions" — D-D reactions for short — ^are
produced by applying high voltage to deuterons (emerging from a
discharge-tube containing heavy hydrogen, in which some of the atoms
are divested of their electrons and the nuclei are left bare) and then
directing them across a vacuum against a target containing other
deuterons. (The target may be some solid compound of heavy
RADIOACTIVITY— ARTIFICIAL AND NATURAL 303
hydrogen, such as ice in which plenty of the hydrogen atoms belong
to the isotope H-; or it may be gaseous heavy hydrogen). The high
voltage is required, so that the impinging deuterons may override the
electrostatic repulsion between the positive charges which they bear
and the positive charges of the deuterons waiting in the target, and
come into contact with these last. Generally in transmutation,
"high voltage" signifies volts by the millions. These particular
reactions are, however, among the easiest to produce, and with less
than a hundred thousand volts it is quite possible to liberate neutrons
at such a rate that their peculiar qualities can be well studied. (One
reaction indeed has been detectably produced at 8000 volts, a figure
so low that it arouses speculation as to what the course of physics might
have been if the second isotope of hydrogen had been discovered say
thirty years ago.)
@ PROTON Q NEUTRON @0 DEUTERON
®o + ©o — ®oo + ©
©o + ®o — - ©©o + o
Fig. 5 — Scheme of the deuteron-deuteron reactions.
To explain what is actually observed to happen, I ask the listener
to imagine the deuteron as a composite of a proton and a neutron, as it
is exhibited in Fig. 5. With this image in mind, one might well expect
that when deuterons are hurled with great energy and speed against
a plate of matter containing massive nuclei — -lead, for instance— they
would be broken in two. This has been sought for but apparently
does not happen, showing that we must keep our imaginations under
continual check by experiment. What does happen is displayed, for
the impacts of deuteron against deuteron, in Fig. 5. It seems that
one deuteron is after all broken in two, but only under the condition
that either its component proton or its component neutron adheres
to the other. Another metaphor: one deuteron snatches either the
proton or the neutron away from the other, leaving the abandoned
neutron or proton to go free. Both of these descriptions are too
figurative, but what is certain is this: from the scene of such impacts,
particles of all the four kinds shown to the right of the arrows in Fig.
5 are observed to be proceeding. The labels show (what should al-
ready be obvious) that the newborn particles of mass 3 are isotopes
304
BELL SYSTEM TECHNICAL JOURNAL
of hydrogen or helium, according as they contain two neutrons and one
proton or two protons and one neutron. I now show pictures to
support these statements.
In Fig. 6 the apparatus is shown in a sketch : the cloud-chamber or
expansion-chamber of C. T. R. Wilson, being the hollow cylinder which
is shown below in axial section, its top being a glass plate and its
bottom a piston-head which can be pulled very suddenly downward
ONCOMING I
DEUTERONSI
k\S\\\\\\\\\1 KWWWWWM
LAMP^
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MICA WINDOWS
K SUPPORTED
ON GRID
Fig. 6 — Expansion-chamber arranged for delecting transmutation by deuterons.
by mechanism. Ordinarily the chamber is filled with moist but dust-
less air; when the piston-head suddenly drops, the air and the water-
vapor are sharply cooled by expansion, and the vapor condenses in
droplets upon whatever ions may be floating in it. The side-tube
which enters the chamber from above is evacuated; through it come
the impinging deuterons, to make their impacts upon the target at the
knob-like closed end of the tube. The wall of the tube, thin as it
may be made, is too thick to allow the deuterons to emerge into the
air of the chamber. One might well expect that a fortiori, any new
particles born out of the transmutation would be too slow-moving to
RADIOACTIVITY— ARTIFICIAL AND NATURAL
305
pierce the wall; but many of these particles are much more energetic
than the impinging deuterons themselves, for they draw upon a
reserve of energy stored up in the nuclei.^ They shoot through the wall
into the air of the cloud-chamber itself, and if they are charged, they
make long trails of ions along their paths. The expansion is then
produced and the water-vapor, condensing upon these ions, makes
trails of droplets which are the paths made visible.
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Fig. 7 — Tracks of a proton and a H^ nucleus resulting from one of the deuteron-
deuteron reactions. (P. I. Dee, Cavendish Laboratory; Proceedings of the Royal
Society.)
Figure 7 exhibits two of these visible paths of "tracks," made by
particles which sprang from the scene of the transmutation (inside the
knob) in practically opposite directions but with very different pene-
trative powers, since one of the tracks is seen to be much longer than
the other. The long one is the track of a proton, the short one is
that of a H^ nucleus which is a deuteron augmented by a captured
neuteron ; this picture shows a single example of the upper reaction of
Fig. 5. How can physicists be sure that these tracks are due to the
nuclei which I have named? This question is far too deep to be an-
swered in this place, and I can only assure the listener that while such
pictures by themselves cannot suffice for the proof, an unassailable
" In the language employed in chemistry, these are "exothermic" reactions.
306
BELL SYSTEM TECHNICAL JOURNAL
proof can be and has been given by other and electrical methods of
observing these newborn particles.
But how about the lower reaction of Fig. 6 — ^the one which really
concerns us, since all this digression is designed chiefly to exhibit the
origin of free neutrons? In Fig. 7 no track appears which can be
attributed to either a He^ nucleus or a neutron; and no such tracks
appear in other similar pictures. The absence of He^ is, however,
due to a simple cause; these nuclei are born with insufficient energy
to traverse the wall of the tube. To observe their tracks it is necessary
to suppress the tube-end, to fill the expansion-chamber with heavy
Fig. 8 — Tracks of protons, H^ nuclei and He^ nuclei resulting from the two
deuteron-deuteron reactions. (P. I. Dee and C. W. Gilbert, Cavendish Laboratory;
Proceedings of the Royal Society.)
hydrogen in the gaseous form, and to project the deuterons directly
into it. When this is done, all of the region of the gas which the
impinging deuterons can reach becomes completely filled with the
ions formed along their many tracks, and appears as a flare on the
photograph (Fig. 8). Out of the flare project the tracks of the new-
born nuclei. Those which stretch clear across the picture are in part
those of protons, in part those of H^ nuclei born from the first reaction.
But in addition one sees a number of short tracks which terminate
not far from the edge of the flare itself. These are the tracks of He^
nuclei — not merely guessed, but proved, to be such.
RADIOACTIVITY— ARTIFICIAL AND NATURAL 307
Where, however, are the tracks of the neutrons? They are not seen
upon this picture, nor in any; for neutrons make no tracks. Neutrons
bear no electric charge, and hence they do not ionize the molecules
of the air or any gas as they go through, for ionization is effected only
by electrical forces which can tear electrons out of their places in
molecules. Not making ions, they afford no footholds whereby the
water-vapor can condense and mark their passage. The expansion-
chamber is frustrated ; and worse yet, so are the electrical devices which
serve for detecting charged particles like fast protons or fast electrons,
since they, too, depend on the ions which these can make. The neutron
indeed might slip through all of our apparatus completely undetected,
were it not liable to make collisions with nuclei so sharp and sudden
F'g- 9 — Track of a proton recoiling from the impact of a neutron. (I. Curie and
F. Joliot, Institut du Radium; Journal de Physique.)
that they might be compared with impacts of one billiard-ball upon
another. Comparisons with billiard-balls are rife in physics, but
seldom with so much justification. When neutrons are streaming
through a gas, such impacts are suffered by nuclei of occasional atoms
of the gas; and like a struck billiard-ball they recoil, and in recoiling
they are able to make ions and the ions then serve to reveal them.
The track of such a recoiling nucleus, made visible in a cloud-
chamber, is seen in Fig. 9. This was taken soon after the discovery
of the neutron, and at a time when these particles had as yet been
released only by using natural radioactive substances to project
alpha-particles against various targets. Such ways of producing free
neutrons are not very efficient, and accordingly one sees in the whole
expansion-chamber one track and one only (though I must interpolate
that according to our present knowledge, many thousands of neutrons
308
BELL SYSTEM TECHNICAL JOURNAL
must have traversed this chamber without happening to strike any
nucleus). See now the contrast with the present time, as illustrated
by Fig. 10 which shows an expansion-chamber traversed by the
Fig. 10 — Tracks of protons recoiling from impacts of a dense stream of neutrons.
(F. N. D. Kurie, University of California.)
neutrons released when deuterons from a high-voltage machine bom-
bard a target.'^ The machine in question was the famous cyclotron
of E. O. Lawrence; there is no more striking illustration of the powers
7 This was another reaction than the one just mentioned, the target being of
beryllium.
RADIOACTIVITY— ARTIFICIAL AND NATURAL 309
which this instrument has conferred upon the scientific world, over
and above those which radium has already granted.
Our digression now ends, and we return to the artificial radioactive
substances, being now equipped with know^ledge as to how these — or
rather, many of them — ^can be made. As I said earlier, many radio-
active isotopes differ from existing stable isotopes only in possessing
an extra neutron in the nucleus; and this extra neutron can be supplied
to the stable nucleus, combining it and converting it into the radio-
active type. I have just exhibited one way in which the extra neutron
may be, and often is, supplied. In the first-mentioned of the deuteron-
deuteron reactions, a neutron is taken away from one of the deuterons
by the other, which latter is thus converted from H- into H^ There
are many stable isotopes, of many elements, which are able to take
away neutrons from impinging deuterons in this manner; a recent list
gives no fewer than fifty. The resulting nucleus-types are not in
every case radioactive; several are stable, including H-^ itself (at least,
no one has yet discovered evidence that H^ is unstable, though there
are doubts about it). Most however are radioactive. The reactions
in question are known as {d, p) reactions, in allusion to the fact that
deuterons enter the target and protons spring out. One might imagine
that the deuteron consists of a proton leading along a neutron, which
it pushes into the nucleus which it strikes, itself continuing its career
as a free particle.
Neutrons, however, do not have to be escorted into nuclei by
protons; those which are already free, such as the ones which are
released in the second of the D-D reactions, are quite well able to
creep in themselves and make themselves permanently at home.
My use of the verb "creep" is not entirely fanciful, for the slower the
neutrons are moving as they approach a target, the better their chance
of entering its nuclei. Those fresh from their origin in reactions of
transmutation are usually moving much too rapidly to be able to come
to a halt in a nucleus — or to be liable to capture, whichever way of
putting it one may prefer. It is necessary to interpose, between the
source and the target, a block of parafifin or a can of water several
inches thick. If the source consists of a natural radioactive substance
bombarding another element with alpha-particles and thus releasing
free neutrons, the two may be mixed with each other and enclosed
in a capsule which is then embedded in the centre of a parafifin sphere
or immersed in water. As the neutrons make their way out, they
collide again and again with the nuclei of atoms in the parafifin or
the water, and these recoil from the impacts. It is not, however,
their recoiling which is now of importance, but the fact that at every
310 BELL SYSTEM TECHNICAL JOURNAL
such impact the neutron loses some of its energy. The point about
choosing water or paraffin is that they are rich in hydrogen and
consequently full of protons, and the elastic impacts of the neutrons
against these entail a greater average loss of neutron-energy per
collision than do impacts against any other nuclei.^ There is good
reason to believe that most of the emerging neutrons have energies
no greater than those which the atoms of the water or the paraffin
possess by virtue of their thermal agitation. These are the neutrons
which are most effective in converting stable into radioactive nuclei
by letting themselves be captured.
Even yet I have not mentioned all of the ways in which radioactive
substances can be and are being made. Time does not suffice for
commenting on the others, but some are exhibited in Fig. 11, which
24 25 26 27 28 29 30 31
12 Mg O 0>.^0
13 Al
14 Si
15 P -Q
Fig. 11 — Various ways of making the radioactive nucleus AP^
displays all of the stable but only one of the unstable isotopes of the
four elements numbered 12 to 15. This one radioactive type, which
I have tried to make more conspicuous by leaving out the rest, is the
isotope 28 of aluminium — ^one of the few elements, which, very con-
veniently for physicists, has one stable isotope only. The arrows
converging onto the star show the different ways in which AF^ is made.
The two coming from the left signify that it is made by adding a
neutron to the stable isotope Al^''; there are two of them, because the
Al^'^ nucleus can either absorb a slow free neutron or annex the neutron
from an impinging deuteron, whichever it has the opportunity of
doing. The arrow slanting downward from the left signifies that AP^
can be made by bombarding magnesium with alpha-particles; some
of these are absorbed by nuclei of the isotope Mg^^ which thereupon
at once emit protons. The arrow slanting upward from the right
signifies a process in which phosphorus is bombarded by neutrons
(fast ones, in this case!) and some are absorbed by the nuclei P^^
* Energy-transfer between an initially-moving and an initially-stationary elastic
sphere is greatest when the latter is of the same mass as the former, and for protons
and neutrons this equality of mass is realized within 0.1 per cent.
RADIOACTIVITY— ARTIFICIAL AND NATURAL 311
which thereupon eject an alpha-particle apiece. The arrow pointing
straight up signifies a process in which silicon is bombarded by neu-
trons; some are absorbed by nuclei Si-^, which instantly throw out
protons. With no fewer than five ways of making a single radioactive
type at his command, the physicist is in a position of power which
seems all the more remarkable when one recalls that as lately as five
years ago he had not (knowingly) made any radioactive substance by
any way whatever.
Consider now the arrow pointing away from the solitary star in
Fig. 11, and the arrows pointing away from the many stars in Figs. 2
and 3. These signify what is really meant by calling an isotope
"radioactive." A radioactive nucleus is one which spontaneously
changes itself into a nucleus of another element by emitting a charged
particle. (Usually it lasts an appreciable time before it does so, and
this delay is to be mentioned in a complete definition of the word
"radioactive.") The arrows pointing away from the stars will serve
to specify these changes. All in these figures are vertical; every one
of these unstable isotopes transforms itself by emitting a particle of
which the mass is very small (compared to the mass-unit which we
are using) while the charge is + e or — e, according as the transfor-
mation is to the element preceding or to the element following. These
particles are positive and negative electrons. All of the man-made
unstable nuclei are radioactive after this fashion, being electron-
emitters; and so are more than half of those which are found in Nature.
What decides whether it shall be a positive or a negative electron
which a given nucleus- type emits? Physicists cannot explain this as
yet, in any adequate sense of the verb "to explain " ; but we can readily
see the law which governs the choice by examining the pictures. In
Figs. 2 and 3 it will be seen that from each star the arrow points in
whichever sense — ^upward or downward — -it finds a circle to point at.
Becoming completely animistic for the moment, I may say that the
unstable nucleus wants to be stable — knows that one of its two
neighbors, of identical mass-number but greater or lesser charge, is
stable — knows which of the two is stable — and deliberately proceeds
to identify itself with its stable neighbor by emitting an electron of
the necessary sign. Putting the situation more drily: each of these
unstable nucleus-types tends to transform itself into its adjacent
stable isobar. Here "isobar" is a technical term for "nucleus of
the same mass-number," and "adjacent" is a short way of saying
"belonging to the preceding or the following element."
Suppose the star has circles both above and below it, i.e. that both
of the adjacent isobars are stable (and prove themselves so by existing
46
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47
48
Ag
Cd
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312 BELL SYSTEM TECHNICAL JOURNAL
in Nature) : how will the unstable nucleus resolve its dilemma? Such
cases are rare, but not entirely absent. An example appears in Fig. 12.
The elements palladium and cadmium have isobaric isotopes of mass-
number 106, in spite of the fact that their atomic numbers (46 and 48)
are not consecutive; silver, with atomic number 47, lies between.
There is no stable silver isotope 106, but a radioactive one can be and
has been created, and for this the dilemma is posed. It handles the
104 105 106 107 108 109
O
O O
o
Fig. 12 — Illustrating an example of isomers.
dilemma by grasping both horns! electrons of both signs come out of
the radioactive silver. I must say that there is something which
indicates that the nuclei which make one choice may be slightly
different (in mass, for instance) from those which make the other.
It may therefore be well to speak of silver as having two isotopes of
the same mass-number 106, and a word has already been coined:
they are called "isomers" of one another. This does not alter the
fact that where alternative choices exist, both are elected.
On Fig. 2 we notice an arrow which points to a vacancy. No
stable nucleus Be^ is known, though there has been a very diligent
search for it conducted by many ways in many places. Practically
no doubt exists that Be^ bursts of itself into two pieces (two alpha-
particles) almost as soon as it is made. We thus have here an unstable
(radioactive) nucleus— Li^ — -which does not find stability by ejecting
an electron, but instead hastens onward to a completer ruin. In the
lower reaches of the Table of the Elements there are so many stable
isotopes that the unstable ones can almost always turn themselves
by one electron-emission into one or another of these, and such
catastrophes are rare. Among the natural radioactive substances in
the upper reaches of the Table they are common, as I now show in
returning to natural radioactivity for the close of this talk.
Notice again Fig. 4, in which the stars are so many and the circles
so few. If arrows were to be inserted to show the transformations,
they would crisscross into a maze. I have therefore separated the
figure into three: all of the circles, stars and rosettes in itwill be found
RADIOACTIVITY—ARTIFICIAL AND NATURAL
313
in Fig. 13 (which is really a pair of figures, as the caption says) and
Fig. 14.
208 212
216
82
83
Fig. 13 — Part of the thorium series of radioactive nuclei. To obtain the actinium
series, imagine each star and rosette transposed one unit to the left.
210
218
\
1
84 >F '^C' ■-)fA
Fig. 14 — Part of the radium series of radioactive nuclei.
Taking Fig. 13 as it stands, we see five of the stars and one rosette
of Fig. 4 connected by arrows; each star marks a nucleus which
transforms itself by emission of a particle into the one next following
along the arrow-chain. The rosette stands for a stable nucleus, and
would be replaced by a circle were it not distinguished by being the
terminus of such a chain. These five radioactive isotopes and one
terminal nucleus-type belong to the "thorium series," and are known
as thorium A, thorium B, thorium C and so on, according to the
letters which adjoin their stars. This is an unlucky bit of terminology,
for it suggests that all are isotopes of the same element, which is
clearly far from the truth.
Taking Fig. 13 again and imagining each star moved by one unit
to the left (so that e.g. the star A goes to 217), we now are thinking
of five more stars and another rosette of Fig. 4, duly connected by
arrows. These constitute (a part of) the "actinium series" and are
known as actinium A, actinium B and so forth.
Taking Fig. 14 as it stands, we find ourselves confronted by eight
more of the stars and the last rosette of Fig. 4, connected by arrows.
These constitute a part of the "radium series" and are known as
k
314 BELL SYSTEM TECHNICAL JOURNAL
radium A, radium B and so on. The "so on" covers more than it
did in the other two cases, this chain continuing to the terminus here
marked as radium G, though usually known by a different name.
Surveying the scene of these massive radioactive nuclei, one is
struck by the fact that not all of the arrows are vertical. Many are
slanting, and by their slant and their length they show that they
represent the emission of alpha-particles. It is a feature of some (not
of all) of the unstable nuclei of mass-numbers greater than 200, that
they strive toward stability by emitting these. For this feature we
should be very grateful, since it was by the use of alpha-particles from
natural radioactive bodies that Rutherford achieved the first of
transmutations; though physicists now can transmute without their
aid, no one can guess how long they would have waited without trying
had they not had that encouragement. Vertical arrows also are seen,
but again there is a contrast to the lighter isotopes; all of the electrons
emitted by radioactive nuclei of mass-numbers beyond 200, or by
natural radioactive isotopes of whatever mass, are negative. But for
the fact that positive electrons had been observed among the cosmic
rays in 1932, they would have been discovered along with the first
examples of artificial radioactive isotopes in 1934, and what a sensation
that would have been !
More than by anything else, probably, one is impressed by the
concatenation of these radioactive nuclei. A long journey to stability
lies ahead of thorium A and actinium A, a longer one still ahead of
radium A; but the total lengths of the journeys are greater yet, for
they begin farther back. In Fig. 15 we behold the three series of
radioactive isotopes in their entirety, and it is seen that the three
"A-products," as they are called, are midway in the evolution and
not at its beginning. The manner of drawing of this figure is changed
from the preceding, atomic number being laid off along the horizontal
axis and mass-number along the vertical; also, crosses and circles and
dots are used to mark the members of the actinium, radium and
thorium series respectively, and have no bearing on stability or
instability. The actinium series should lie lower than it is drawn,
with its terminus AcD lying midway between ThD and RaG; the
mistake is incurred so as to diminish the overlappings which would
otherwise confuse the picture.
Except for a few created in the last three years by transmutation,
every known nucleus-type of mass-number greater than 209 and
atomic number greater than 83 (as well as a few of slightly lower
values) is found in Fig. 15. It appears that 83 and 209 are critical
values of nuclear charge and mass, beyond which the constituents of
RADIOACTIVITY—ARTIFICIAL AND NATURAL
315
nuclei — ^neutrons and protons, presumably, and whatever others there
may be — cannot unite into stable systems.^ All of these nuclei
beyond 209 which are found in Nature are seeking for stability by the
emission of particles, but never finding it until they have emitted
80 81 62 83 84 85 86 87 88 89 90 91 92
ATOMIC NUMBER
Fig. 15 — The three series of radioactive substances.
sufficiently many to convert themselves (or rather, the residues of
themselves) into one or another of the three isotopes of element 82 — ■
lead — which are marked by rosettes in Fig. 4. For an obvious reason
these three are called thorium lead, actinium lead and radium lead.
"Ordinary" lead as it comes from most mines is a mixture of many
isotopes, but the lead which is found in close association with thorium
or with uranium proves its origin from vanished atoms of these metals
' I recall from an earlier footnote that some of these very heavy nuclei (notably
thorium 232 and uranium 238) are so very long-lasting that "unstable," while strictly
correct, seems much too strong a word for them.
316 BELL SYSTEAI TECHNICAL JOURNAL
by being preponderantly the isotope 208 or the isotope 206 as the case
may be.
I pause to mention, in justice to Rutherford, that it was he who
proved by study of some of these natural radioactive bodies that each
is transforming itself into a different element; also it was his associate
Soddy who by similar studies was led to distinguish the first-to-be-
recognized isotopes, that is, different radioactive forms of one and the
same element. The way for making such diagrams as Figs. 2 and 3
and 4 was prepared before 1914 by these two men, though some of
the facts embodied in Fig. 4 were not then available because of want
of knowledge of atomic numbers, and all of the knowledge embodied
in Figs. 2 and 3 was non-existent because no radioactive isotopes of
these elements had yet been created and nobody knew as yet how to
distinguish their stable isotopes. Also it should be mentioned that
only the extraordinary potency of radioactive substances in affecting
our instruments of measure enables the physicist or the chemist to
recognize the element to which a radioactive isotope belongs, nay even
to detect its presence. With radium and a few others, it has been
possible to amass enough of the substance to see and to weigh; with
the great majority of natural and with the totality of artificial radio-
active isotopes, nothing of the sort has even been approached, and we
should still be unacquainted '" with them if they had been stable.
Three examples of transmutation which occur in these upper ranges
of the Periodic Table deserve to be recorded in even so brief a report.
In Fig. 14, notice the circle in row 83 and column 209 which (as I
earlier said) represents the highest stable nucleus — ^bismuth 209.
Radium E, represented by the star to its right, is clearly bismuth 210;
by the testimony of its mass-number and atomic number, it differs
from stable bismuth nuclei by the possession of an extra neutron.
If bismuth should be bombarded by neutrons, either free or bound
into deuterons, would it be transformed into radium E? Livingood
at Berkeley did bombard ordinary bismuth with very energetic deu-
terons, and did succeed in producing a radioactive substance which
agreed with radium E not only in emitting negative electrons, but also
in converting itself into a substance emitting alpha-particles, and the
agreement extended to details of the emission. No doubt exists
that he was making radium E out of bismuth 209 by enabling deuterons
to transfer their constituent neutrons to this latter, just as H^ is made
from H^ in the first of the deuteron-deuteron reactions.
1" This statement should be quaHfied slightly, for some of the artificial radioactive
nuclei spring from reactions of transmutation which are so well understood that the
observer could justifiably infer the existence of the nuclei in question even if he did
not observe them.
RADIOACTIVITY— ARTIFICIAL AND NATURAL 317
In F'ig. 15. notice that all of the members of the thorium series have
mass-numbers divisible by 4, or equal to 4w with various integer values
given to n. This is accordingly called the "4w series," and one
readily sees what is meant by calling the radium and the actinium
series by the names of "4n + 2" and " 4w + 3" series respectively.
One begins at once to wonder whether there is not a "4n + 1 " series.
Such a series was long sought after in vain, and no member of it has
yet been discovered in Nature; but in the laboratory of the Curies in
Paris thorium has lately been strongly bombarded by neutrons, and a
new sequence of radioactive bodies has thus been engendered which
has already been followed through several steps, and is in all probability
the series so long missing.
As to the remaining feat — the creation of elements beyond uranium
— it is now beyond doubt. Fermi and his school at Rome, Hahn and
Meitner in Berlin, Curie and Joliot in Paris have all borne witness to
it. In one way it seems the most romantic of all the feats of transmu-
tation, for Nature had apparently set 92 as the limit of nuclear charge,
and now man has transgressed it. The process is begun by exposing
uranium to bombardment by streams of neutrons. It appears that
when a uranium nucleus has captured a neutron, it finds itself not
strongly enough charged to hold together, and proceeds to emit one
negative electron after another in its search for stability. Each
emission transfers the nucleus to an element one step higher, without
afifecting its mass-number; and the authorities agree that there are at
least four consecutive emissions, after the last of which the atomic
number is 96! This addition of four new elements to the Periodic
Table opens a new field to chemists, one which they can scarcely
have expected ever to be able to enter. The four have no proper
names as yet, a curious circumstance in view of the fact that dis-
coverers of new elements have thus far been in great haste (sometimes
too great haste) to name them. Mendeleieff long ago used to denote
an expected but undiscovered element by prefixing "eka" to the name
of the element just above the vacant place in the Periodic Table;
these new four are sometimes called eka-rhenium, eka-osmium, eka-
iridium and eka-platinum, but on looking at such words one is inclined
to prefer the atomic numbers.
Now to summarize. The world as we knew it before the days of
transmutation was constructed out of some two hundred and fifty
kinds of atoms, each consisting of a nucleus surrounded by a family of
electrons. Of these 250 kinds of nuclei the great majority were stable
and perpetual, but some forty were unstable — ^doomed to perish in
due time, by ejecting either alpha-particles or negative electrons.
318 BELL SYSTEM TECHNICAL JOURNAL
These were the natural radioactive bodies. To these forty kinds of
radioactive nuclei already found in Nature, physicists have added in
a scant four years no fewer than two hundred and twenty more by
the art of transmutation. Every chemical element which is known to
exist at all, with the sole exception of hydrogen, has at least one
radioactive type of nucleus or isotope, and many have more than one.
These man-made radioactive nuclei are often made simply by adding
neutrons to nuclei which already exist and are stable. There are,
however, other and more complicated processes, in which neutrons or
protons or deuterons or alpha-particles impinge on nuclei and seem to
enter them, and other particles leap out. Many radioactive bodies
have already been made in two or three different ways, some in as
many as five.
Few things are riskier than to suggest a limitation either on the
scope of Nature or on the possibilities of science, and many a scientist
is remembered chiefly for such a suggestion which later the course of
events proved foolish. Yet there are circumstances in this case which
give some ground for suspecting that already we may know nearly
all of the stable and may have created nearly all of the radioactive
nucleus-types. Several hundreds of types have now been made by
the art of transmutation, but of them nearly all which seem to be
stable are not new, and nearly all which are new are radioactive.
This implies that the earth has already been stocked with almost all
the stable nucleus-varieties, but not necessarily that we have yet come
near to making all of the possible radioactive kinds. There are how-
ever reasons for believing that most of the remaining types have so
little durability, that even if they were to be made they would not
last long enough to be identified as radioactive. Nature probably
has come quite close to building all the imperishable forms, we possibly
almost as close to creating all of those which are capable of a little
but not a perpetual life. Perhaps it is fitting that people who are not
immortal should not be able to construct new elements which are
immortal; but we at least can rejoice in having diversified the scene
of the world with a surprising number of new substances which are
none the less remarkable for being transitory.
Abstracts of Technical Articles from Bell System Sources
Protection Features for the Joint Use of Wood Poles } J. O'R. Cole-
man and A. H. Schirmer. The paper reviews the historical develop-
ment of joint use and the general results to date of studies of protective
problems of lower and higher voltage joint use. The safety features
are reviewed from the standpoint of (1) subscribers' premises, (2) em-
ployees, and (3) telephone plant. Characteristics of equipment of
power and telephone plant as far as they relate to this problem are
given. The various factors which determine magnitude and duration
of the current and voltage in the telephone plant resulting from a con-
tact with power conductors are discussed. Improved methods for
obtaining safety under various conditions, where higher voltage joint
use is found to be the best over-all solution, are described.
High-Speed Motion Picture Photography Applied to Design of Tele-
phone Apparatus? W. Herriott. High-speed motion pictures are
employed at Bell Telephone Laboratories as a visual aid in the study
of problems associated with the design, manufacture, and testing of
telephone apparatus. A new high-speed camera of the optical com-
pensator type operating at 4000 pictures per second is described, and
its application to the study of problems associated with telephone
apparatus is discussed.
Mass Ratio of the Carbon Isotopes from the Spectrum of CN.^ F. A.
Jenkins and Dean E. Wooldridge. With a source containing carbon
enriched about ten times in C^^, the violet CN bands have been photo-
graphed with a dispersion of 0.63A/mm. Measurements are given of
the lines of low rotational quantum number in the 0,0, 0,1 and 0,2
bands of C^^ N"^"*, as well as of C^^ N^"*. The vibrational constants of the
normal states of both molecules are accurately determined, and give a
value of the isotope mass coefficient p = oj^/w, of 0.97898 ± 0.00002,
corresponding to a mass for C^^ of 13.0088. This is in essential agree-
ment with the mass-spectrograph value, and it is shown that the finer
corrections to the isotope effect are negligible in this case.
» Elec. Engg., March 1938.
2 Jour. S. M. P. E., January 1938.
3 Phys. Rev., January 15, 1938.
319
320 BELL SYSTEM TECHNICAL JOURNAL
Composition and Structure of Hevea Latex} A. R. Kemp. Data and
present views relating to the composition and structure of the latex
particles are presented. The number of particles in one gram of 40
per cent latex was calculated to be 7.4 X lO^^ q^i the basis that they
have an average particle mass of 0.054 X lO^^^ gram (from the micro-
scopic data of F. F. Lucas, page 146).
A study was made of the efifect of several factors on the water content
of rubber in pressed coagulum from fresh and treated latices. The
average value for the water of retention of rubber coagula from fresh
latex was found to be about 1 1 per cent, increasing to about 22 per cent
in the case of old latex and deproteinized rubber from alkali-treated
latex. This water appears to be held mechanically in the colloid hydro-
carbon structure of the latex particles.
The particle structure of sheet rubber is discussed and it is suggested
that plasticization by milling involves the conversion of the gel hydro-
carbon shell on the rubber particles to sol rubber through oxidation.
Ultraviolet Microscopy of Hevea Rubber Latex.^ Francis F. Lucas.
Samples of bulk rubber latex received in sealed cans from two sources
have been investigated by means of the ultraviolet microscope. The
advantages of the ultraviolet microscope are (a) an enormous increase
in resolving power, (b) selective absorption of the ultraviolet light by
many substances, and (c) the ability to section optically very small
objects suited to the purpose.
Brief descriptions of the apparatus and technic are given. Artifacts
have been minimized in the preparation of the slides. A multitude of
particles bordering on colloidal dimensions have been clearly resolved.
Particle size measurements, including complete tabular data and a
particle size distribution curve for each specimen, are given. Ap-
proximately 90 per cent of the particles are 0.50 micron or less in
diameter. The shape of the latex particle appears predominantly
spherical, although elongated particles and irregular shaped particles
are found. Optical sections in some cases show these to be groups of
particles; two particles may coalesce to form one. Many of the
smaller particles appear to lose their electrical charges and become
attached to larger particles. Possible effects of ultraviolet radiation
are discussed.
Dielectric Losses in Polar Liquids and Solids.^ S. O. Morgan.
Dielectric loss is the energy dissipated as heat in a dielectric when it is
in an electric field. Losses due to dipoles represent only one of a num-
* Indus, and Engg. Chem., February 1938.
6 Indus, and Engg. Chem., February 1938.
« Indus, and Engg. Chem., March 1938.
\
ABSTRACTS OF TECHNICAL ARTICLES 321
ber of possible means by which energy may be dissipated in a dielectric;
there may be losses due to free ions and also due to dielectric polariza-
tions other than dipole polarizations. However, in many materials
dipoles are an important source of loss; it is the purpose of this paper
to consider some of the typical cases of dipole loss and to point out some
of the relations between chemical composition and dipole loss which
follow from the recent experimental and theoretical study of dielectric
behavior.
Order-Disorder Transformations in Alloys.'' Foster C. Nix and
William Shockley. An extensive resume of a subject which is
becoming of increasing interest to physicists and metallurgists. The
article divides into two parts, forty pages being devoted to the theories
of the order-disorder phenomenon, and twenty pages to experimental
studies of superstructures.
Thyratrons for Grid-Controlled Rectifier Service.^ G. H. Rockwood.
It is common knowledge that the output voltage of a rectifier fluctuates
with changes in load current and supply line voltage. Frequently
these fluctuations are so large that means must be used to correct them.
This is particularly true when the rectifier feeds a load having a high
back electromotive force and a small resistance, such as a storage bat-
tery. The facility with which the output voltage may be controlled
by the use of thyratrons as the rectifying element has encouraged the
design of tubes especially suited to this purpose. There is available a
variety of circuits such that the output voltage of a rectifier may be
made to obey any desired law. The successful application of these
circuits depends upon the degree of reliability of the thyratron tubes
used in them. To be most successful the tubes must possess certain
characteristics. This paper gives a brief review of the operation of
grid-controlled rectifier circuits, discusses the requirements which such
circuits impose on the tube characteristics, and describes a particular
type of thyratron with mercury-plus-argon filling which has proved
especially useful in such rectifiers.
Progress in Non-ferrous Metals and Alloys During the Past Few
Years.^ Earle E. Schumacher and Alexander G. Souden. The
purpose of this review is to present the more important advances in
the non-ferrous field during the past few years, the topics discussed
being classified broadly as fundamental and practical. The former
' Reviews of Modern Physics, January 1938.
* Trans, the Electrochemical Society, Vol. LXXII, 1937, pp. 213-224.
* Mining and Metallurgy — Institute of Metals Division, January 1938.
322 BELL SYSTEM TECHNICAL JOURNAL
includes those studies that have done most toward developing the basic
science of metals and alloys, and the latter includes technical develop-
ments and applications.
Theory of Order for the Copper Gold Alloy System}^ W. Shockley.
The theory of order and disorder, in the form used by Bragg and
Williams, is extended to arbitrary composition of the constituent ele-
ments. The work is based upon the nearest neighbor interaction
assumption of Bethe and the connection between the Bethe and Bragg-
Williams theory is shown. In order to extend the Bragg-Williams
theory to compositions other than 25 and 50 atomic per cent, new
definitions of order are developed. The results are presented in terms
of phase diagrams and curves showing energy vs. temperature, specific
heat vs. temperature and state order vs. temperature. These results
are of importance in giving a general picture of the order-disorder
transformation for a wide composition range. They are not in detailed
accord with experiment due to the rather idealized picture underlying
the nearest neighbor assumption.
A Theory of Noise for Electron Multipliers}^ W. Shockley and
J. R. Pierce. The noise in secondary-emission electron multipliers is
considered from a theoretical viewpoint. The noise properties of a
stage are correlated with its secondary-emission properties: the mean
value m and mean-square deviation 5^ of the number of secondaries per
primary. If /pA/ and /^a/ denote the mean-square noise current
lying in the frequency band A/ in the primary- and secondary-electron
currents, then 7sa/ = m^ ^pa/ + h'^lel^^ where /p is primary direct
current. This result is applied to many-stage multipliers. For n
similar stages 7,"I7 = MlPpA/ + h'^{_M{M - \)\m{jn - l)]2e/^ where
M = m" is the over-all gain of the multiplier.
Wave Guides for Electrical Transmission}"^ G. C. Southworth.
The transmission of electric power at extremely high frequencies
through rods or "wires" of dielectric and through metal tubes, without
the usual return conductor, was predicted mathematically many years
ago. Recently experiments have confirmed this theory. Wave guides
offer the possibility of transmitting very wide frequency bands and
consequently extremely large numbers of speech channels without the
high attenuations encountered in radio; in fact, constantly decreasing
attenuation with increasing frequency is predicted for one type of wave.
10 Jour. Chemical Physics, March 1938.
11 Proc. I. R. E., March 1938.
12 Elec. Engg., March 1938.
ABSTRACTS OF TECHNICAL ARTICLES 323
Some of the properties of the waves and the apparatus used In studying
them are described in this article.
Recent Development in Hill and Dale Recorders P L. Vieth and C. F.
WiEBUSCH. A new sound-on-disk recorder has been developed in
which is used the principle of feeding part of the output of the system
back to the input of the associated driving amplifier in properly
controlled relationship. The use of this principle, which is widely used
in feedback amplifiers, replaces the usual practice of providing dis-
sipative elements for the control of an electrically driven vibrating
system. Heretofore no practical application of feedback to electro-
mechanical systems has been made, possibly because the requirements
for stable operation of such systems are difficult of achievement.
Through recent developments these requirements have been satisfac-
torily met. The new recorder is capable of recording on wax or direct-
recording material without appreciable effect upon its characteristics,
which include uniform response from 30 to 12,000 cps. and exceptional
freedom from distortion. The recorder is extremely simple and affords
easy means for field calibration from the feedback element, whose
output is in direct proportion to the stylus velocity. These means also
make available a monitoring voltage which, properly amplified, gives a
precise aural picture of the stylus behavior during recording.
Internal Friction in Solids — ///. Experimental Demonstration of
Thermoelastic Internal Friction}^ C. Zener, W. Otis and R. Nuc-
kolls. In order to demonstrate the presence of thermoelastic internal
friction, the authors measured the internal friction of a copper reed
over a wide frequency range (50 to 4000 cycles/sec). They obtained a
maximum precisely at the predicted frequency. The observed varia-
tion of internal friction with frequency proves that, over a wide
frequency range, the internal friction due to the flow of heat back and
forth across a reed is of a larger order of magnitude than that due to all
other causes. Independent experiments of Bennewitz and Rotger on
wires of silver, aluminum, brass, steel, and glass are shown to furnish
an equally striking demonstration of thermoelastic internal friction.
13 Jour. S. M. P. E., January 1938.
^*Phys. Rev., January 1, 1938.
Contributors to this Issue
H. W. Bode, A. B., Ohio State University, 1924; M.A., 1926; Ph.D.,
Columbia University, 1935. Bell Telephone Laboratories, 1926-.
Dr. Bode has been engaged in the study of transmission networks, such
as wave filters, attenuation equalizers, and phase correctors.
James A. Carr, B.S. in Electrical Engineering, Virginia Polytechnic
Institute, 1919; Instructor in Electrical Engineering at Massachusetts
Institute of Technology, 1920. American Telephone and Telegraph
Company, Development and Research Department, 1921 27; Bell
Telephone Laboratories, 1927-. Mr. Carr has been engaged prin-
cipally in outside plant development work.
Karl K. Darrow, B.S., University of Chicago, 1911 ; University of
Paris, 1911-12; University of Berlirt, 1912; Ph.D., University of
Chicago, 1917. Western Electric Company, 1917-25; Bell Telephone
Laboratories, 1925-. Dr. Darrow has been engaged largely in writing
on various fields of physics and the allied sciences.
C. O. Gibbon, B.S., Purdue University, 1914; M.S., Massachusetts
Institute of Technology, 1917; M.E.E., Harvard University, 1917;
Electrical Engineering Laboratory Assistant, Massachusetts Institute
of Technology and Harvard University, 1917-18. American Tele-
phone and Telegraph Company, Department of Operation and En-
gineering, 192 1-. First engaged in work on telegraph matters, Mr.
Gibbon has since 1925 been occupied principally with problems per-
taining to local transmission and in the preparation of engineering
data for the transmission design of the exchange plant.
F. V. Haskell, B.S. in Electrical Engineering, Worcester Poly-
technic Institute, 1926. New York Edison Company, 1926-29.
Bell Telephone Laboratories, 1929-. Mr. Haskell has been engaged
principally in aerial systems work.
Arthur W. Horton, Jr., A.B., Princeton University, 1920; E.E.,
Princeton University, 1922. Western Electric Company, Engineering
Department, 1922-25; Bell Telephone Laboratories, 1925-. Mr.
Horton has been engaged in work relating to the application of voice
operated devices in the toll telephone plant and in studies of toll
transmission.
324
CONTRIBUTORS TO THIS ISSUE 325
O. J. Murphy, B.S. in ELlectrical Engineering, University of Texas,
1927; Columbia l^niversity, 1928-31. Bell Telephone Laboratories,
1927 . Mr. Murphy has been principally engaged in studies of the
effects of transmission delay and voice operated devices on toll tele-
phone circuits.
A. C. NoRwiNE, A.B., University of Missouri, 1923; B.S. in Elec-
trical Engineering, 1924; E.E., 1925. Bell Telephone Laboratories,
1925-. Mr. Norwine has been principally engaged in studies of the
effects of transmission delay and voice operated devices on toll tele-
phone circuits.
I
\yj . » . Ti • • » • r»*^ «j.
VOLUME XVn JULY, 1938 NUMBER 3
THE BELL SYSTEM
TECMSflCAL JOURNAL
DEVOTED TO THE SCIENTIFIC AND ENGINEERING ASPECTS
OF ELECTRICAL COMMUNICATION
Hertz, the Discoverer of Electric Waves — Julian Blanchard , 327
Instruments for the New Telephone Sets — W, C. Jones . , 338
Transmission Features of the New Telephone Sets
—il. H, Inglis 358
Spectrochemical Analysis in Communication Research
—Beverly L. Clarke and A. E. Ruehle 381
High Speed Motion Picture Photography — W. Herriott . . . 393
An Optical Harmonic Analyzer — H. C. Montgomery .... 406
Magnetic Shielding of Transformers at Audio Frequencies
—W. G, Gustafson 416
Coaxial Cable System for Television Transmission
— M. E. Strieby 438
Stabilized Feedback Oscillators — G. H, Stevenson .... 458
The Discovery of Electron Waves — C. J. Davisson .... 475
Abstracts of Technical Papers 483
Contributors to this Issue 486
AMERICAN TELEPHONE AND TELEGRAPH COMPANY
NEW YORK
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EDITORIAL BOARD
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Copyright, 1938
American Telephone and Telegraph Company
PRINTED IN U. 8. A.
HEINRICH RUDOLPH HERTZ
1857-1894
The Bell System Technical Journal
Vol. XVII July, 1938 No. 3
Hertz, the Discoverer of Electric Waves *
By JULIAN BLANCHARD
FIFTY years have passed since those memorable researches of the
young German physicist, Heinrich Rudolph Hertz, which have
come to be regarded as the starting point of radio. For it was he who
first detected, and measured, electromagnetic waves in space — waves
which had been predicted, it is true, but which had never before been
observed. It is not to be claimed, of course, that the radio art would
have failed to be born were it not for his genius, for we know that almost
simultaneously the experiments of Lodge in England were pointing with
certainty to the same discoveries, and the speculations of others were
revolving around the possibility of generating electric waves. Yet it
was the remarkably clear vision of Hertz, combined with his consum-
mate persistence and skill, that won for him the prize and justly
enshrined his name among the immortal men of science.
So, upon this golden anniversary of the opening of a new epoch in
the realm of communication, it is fitting that we pause to do honor
to his memory and to consider anew the significance of his great
accomplishment.
The formal facts of Hertz's biography can be set down very quickly.
He was born at Hamburg on February 22, 1857, his father an attorney,
belonging to a family of successful merchants, his mother the daughter
of a doctor of medicine, and the descendant of a long line of Lutheran
ministers- — all of cultural tastes and attainments on both sides. At
the age of twenty he went off to school at Munich, after a rather
unorthodox preparatory training, supposedly to pursue an engineering
career, but he was torn between this resolve and his natural inclination
for the study of pure science. Soon after reaching Munich he felt
compelled to put the matter before his parents, to whom he frequently
and confidingly wrote concerning his plans and his work. In a long
letter written in November, 1877, he said, "I really feel ashamed to
say it, but I must: now at the last moment I want to change all my
* Published in Proc. I.R.E., May, 1938, Vol. 26, No. 5.
327
328 BELL SYSTEM TECHNICAL JOURNAL
plans and return to the study of natural science. I feel that the time
has come for me to decide either to devote myself to this entirely or
else to say good-bye to it; for if I give up too much time to science in
the future it will end in neglecting my professional studies and becom-
ing a second-rate engineer. Only recently, in arranging my plan of
studies, have I clearly seen this — so clearly that I can no longer feel
any doubt about it. . . ." And then follows a lofty and appealing
presentation of the reasons for his choice. Concluding, he wrote:
"And so I ask you, dear father, for your decision rather than for your
advice; for it isn't advice that I need, and there is scarcely time for it
now. If you will allow me to study natural science I shall take it as a
great kindness on your part, and whatever diligence and love can do in
the matter, that they shall do. I believe this will be your decision, for
you have never put a stone in my path, and I think you have often
looked with pleasure on my scientific studies ..."
Matters were arranged as he wished and he joyfully pursued his
studies at the University and at the Technical Institute, working hard
on mathematics and mechanics, and spending much time in the physical
laboratory. In October, 1878, he went to Berlin and became there a
student under the mighty giants, von Helmholtz and Kirchhoff ; writing
to his parents in November, "I am now thoroughly happy, and could
not wish things better." In 1880 he gained his doctorate, and in
October of that year was appointed assistant to Helmholtz, which
delightful and stimulating association continued until Easter of 1883.
He then went to the University of Kiel to become lecturer in theoretical
physics, and here he first began to reflect seriously upon Maxwell's
electromagnetic theory of light.
He was soon promoted again, and at Easter, 1885, he became pro-
fessor of experimental physics at the Technical High School in
Karlsruhe Here, in 1886, at the age of twenty-nine, he married
Elizabeth Doll, daughter of the professor of geodesy in the same
institution, and their home became a congenial meeting place for their
many cultured associates. It was at Karlsruhe also that he began his
researches on electric waves. Before they were finished he was called
in 1889 to succeed the celebrated Clausius at the University of Bonn,
thus at the age of thirty-two having arrived at a position in the
academic world not ordinarily to be achieved until much later in life.
He soon thereafter relinquished to others the further exploration of
the great new territory of electric waves he had opened up and re-
turned to some investigations on the discharge of electricity in rarefied
gases, a subject which had interested him while at Berlin. He then
devoted his attention entirely to what proved to be his last work, a
HERTZ, THE DISCOVERER OF ELECTRIC WAVES 329
treatise on mechanics. In the summer of 1892 he suffered a severe
illness which eventually led to chronic blood poisoning, of which he
died, after indescribable suffering, on January 1, 1894. He would have
been just thirty-seven years old the following month.
Although the fame of Hertz rests primarily on his electric wave
researches, these constitute by no means the whole of his work. His
collected papers, edited by the German physicist. Dr. Philipp Lenard,
and admirably translated into English by Professor D. E. Jones and
associates, comprise three volumes. The first consists mainly of his
miscellaneous earlier papers, some twenty-odd titles altogether. One
of these, published in 1884, "On the Relations between Maxwell's
Fundamental Electromagnetic Equations and the Fundamental Equa-
tions of the Opposing Electromagnetics," marked an important step in
the development of Hertz's ideas, and has been called his greatest
contribution to theoretical physics. In it he opposed the old orthodox
theories of electric phenomena based on action at a distance, which
were supported by most of the Continental physicists, and definitely
aligned himself with the followers of Maxwell. This volume also
contains his semipopular Heidelberg lecture of 1889, "On the Relations
between Light and Electricity," giving a general account of his more
recent work. It strikingly illustrates the charm and felicity of style
he could employ in the presentation of a difficult subject, and cannot
fail to be read, and reread, with pleasure and admiration. The volume
ends with a eulogy of Helmholtz on the occasion of his seventieth
birthday, wherein the pupil, in equally graceful language, paid homage
to his beloved and inspiring preceptor. These two papers reveal so
clearly between the lines the manner of man their author was.
The second volume contains the papers on electric waves, which
had been collected by the author himself, as a result of numerous
requests for reprints, and published under the German title, "Unter-
suchungen iiber die Ausbreitung der Elektrischen Kraft," and later in
English as "Electric Waves," with a lucid introduction by Hertz
explaining the motive and significance of each of the separate papers.
One of the first of these describes an important by-product of the
electric wave studies, the discovery of the effect of ultraviolet light
upon an electrical discharge. This discovery itself uncovered a new
wealth of physical problems and the subject became immediately of
great interest to many experimenters, most of whom at the time little
suspected that this subsidiary effect was not the main discovery, or
imagined that electrical science was on the eve of much greater con-
quests. The paper attracted attention to Hertz and aroused a popular
nterest in him, so that everything coming from him thereafter was
330 BELL SYSTEM TECHNICAL JOURNAL
eagerly read. Had it not been for this rather accidental circumstance
the importance of his subsequent work might have gone temporarily
unnoticed, as has happened with some of the greatest discoveries.
The third volume in the series is his book, "Die Prinzipien der
Mechanik," completed with difficulty during his last illness and pub-
lished a few months after his death. A lengthy preface by the vener-
able Helmholtz gives an appreciative sketch of the author's life and
work — this time a sorrowful tribute from master to pupil.
In order to understand and appraise the work of Hertz on electric
waves, it will be needful to review briefly the ideas about light and
electricity that prevailed at his time and before. With regard to light,
Newton's corpuscular theory had given way in the early 1800's to
the wave theory of Young and Fresnel, and long before Hertz's day
the idea of transverse vibrations in a hypothetical elastic-solid medium
called the luminiferous ether had become firmly established. Length
of wave and velocity had been measured many times. The wave
theory accounted satisfactorily for all the known phenomena in optics
and there was no doubt in anybody's mind about its essential correct-
ness, regardless of any difficulties encountered in explaining the nature
of the ether and its relation to matter.
As to electricity and magnetism, the older theories of instantaneous
action at a distance were beginning to weaken with the discoveries,
in the early part of the nineteenth century, of the reactions between
electric currents and magnets, and the phenomenon of induction.
Hitherto there had been no postulation of an intervening medium to
explain the transmission of the force between two charged bodies, and
it was supposed that electricity and magnetism, like gravitation, acted
across empty space in straight lines and instantaneously. In some-
what different dress these ideas were given new life around the middle of
the century, particularly by some of the German physicists. To Fara-
day, however, such views were unacceptable. He wished to get rid of
the idea of action at a distance and in his mind pictured a medium,
along the contiguous molecules or particles of which the force was
propagated. In this medium he visualized "lines of force" emanating
from or terminating upon the electric charges or magnetic poles, acting
like stretched elastic threads, repelling each other sidewise as well as
tending to contract. Thus, in his thinking, attention was focused upon
the insulating medium surrounding a conductor, the "dielectric" as he
called it, for here, he thought, was the real seat of the action. He be-
lieved also that there existed some direct relation between light and
electricity or magnetism. He was ever seeking to find such a relation
and in the course of his many experiments he discovered the rotation
HERTZ, THE DISCOVERER OF ELECTRIC WAVES 331
of the plane of polarization of light by a magnetic field. In his specula-
tions there was one question which was continually presenting itself
to him. Do electric and magnetic forces, like light, require time for
their propagation? Are there waves? But to this question he was
unable to find an answer.
And then came Maxwell, building upon the foundation that Faraday
had laid, translating Faraday's ideas into the language of mathematics,
and making the grand generalization that light and electric waves are
one and the same phenomenon, propagated by the same medium, with
the same velocity, and differing only in wave-length. Like Faraday,
he considered the energy of the electromagnetic field to reside in the
dielectric. He conceived the medium to have properties analogous to
those of an elastic solid, which would spring back to its original state
upon the removal of the straining force. The alteration of the dis-
placement, or "polarization," in the medium was viewed by him as an
electric current, which he called the "displacement current," as dis-
tinguished from the "conduction current" existing in conductors.
From the general equations which he formulated it was shown that only
transverse vibrations (like light) could be propagated in such a medium
and that the velocity of propagation should be equal to the ratio of
the two systems of electrical units, that is, to the number of electro-
static units of electricity contained in one electromagnetic unit. This
ratio had been experimentally determined by Weber and Kohlrausch
(it was later redetermined by Maxwell himself, by a different method),
and the fact that it agreed so very closely with the measured velocity
of light was one of the strongest points in favor of the view that light
waves were identical with the hypothetical electromagnetic waves.
Maxwell's comprehensive theory was first enunciated in his 1865 paper
entitled "A Dynamical Theory of the Electromagnetic Field," and
afterwards elaborated in his great "Treatise on Electricity and Mag-
netism," published in 1873, but for a number of years it was regarded
by many as merely a speculation, by others as probably true, and by
none as conclusively proved. It remained for Hertz to add the cap-
stone to the theory by actually demonstrating for the first time the
existence of electromagnetic waves in space; and furthermore, to show
experimentally that they had all the physical properties of ordinary
light waves.
In 1888, while he was teaching at the Technical High School in
Karlsruhe, Hertz carried out the brilliant experiments which have
made his name famous. These were actually a part of a long series of
experiments which began in 1886, and which came about partly by
accident, and yet were the result of his keen interest in everything
332 ' BELL SYSTEM TECHNICAL JOURNAL
connected with electric oscillations; an interest extending back to
1879, when, at the suggestion of Helmholtz, he had considered tackling
a prize problem proposed by the Berlin Academy of Science aimed at
the proof of a portion of Maxwell's theory, but which he had abandoned
for the reason that oscillations of sufficiently high frequency were not
then available. While using in his lectures at Karlsruhe a pair of
flat ("pancake") coils, called Reiss or Knockenhauer spirals, mounted
adjacent to each other, he was surprised to find how easy it was to
obtain sparks between the terminals of the secondary coil when a small
Leyden jar or even a small induction coil was discharged through the
primary, provided the primary discharge took place across a spark
gap. This, of course, was an indication of an exceptionally strong in-
ductive effect. This observation led to his discovery of a method of
exciting electric disturbances far more rapid than any hitherto known,
such as those of Leyden jars or open induction coils as customarily
used, and having wave-lengths, it turned out, capable of being meas-
ured within the confines of a laboratory. His oscillator was nothing
more than a short metal rod with a spark gap in the middle (sometimes
with metal spheres or plates attached to the ends, resembling a dumb-
bell), the sparking terminals consisting of small knobs or spheres which
were connected to the terminals of a Ruhmkorff induction coil; the
small inductance and capacitance of this simple linear conductor, to-
gether with the proper functioning of the spark gap, accounting for its
success. By such means Hertz obtained wave-lengths from a few
meters down to 30 centimeters, and so began, it is seen, with the " ultra-
short" waves that are again coming into vogue.
Hertz began his experiments with a study of the "induction " about
this exciter. As he commented in his first paper in this series, "On
Very Rapid Oscillations," published in May, 1887, theory had pre-
dicted the possibility of very rapid oscillations in open-wire circuits of
small capacitance, but it could not be predicted from theory whether
they could be produced on such a scale as to admit of their being ob-
served. Hertz not only devised a method of producing such oscilla-
tions, but also discovered a method of detecting them, by their effects
in the surrounding space. His detector consisted merely of a short
length of wire bent in the form of a rectangle or a circle and containing
a micrometer spark gap, across which minute sparks could be seen in a
darkened room; especially if this secondary circuit was in electrical
resonance with the exciter. This exceedingly simple detector was
indeed a capital discovery. Some five years earlier Professor G. F.
Fitzgerald, of Dublin, had suggested "the combination of a vibrating
generating circuit with a resonant receiving circuit ... as one by
HERTZ, THE DISCOVERER OF ELECTRIC WAVES 333
which this very question might be studied." But, as he said after-
wards in speaking of Hertz's work, "I did not see any feasible way of
detecting the induced resonance: I did not anticipate that it could
produce sparks." Concerning this contrivance the following interest-
ing remarks were made by its author in his Heidelberg lecture above
referred to: "The method had to be found by experience, for no amount
of thought could well have enabled one to predict that it would work
satisfactorily. For the sparks are microscopically short, scarcely a
hundredth of a millimeter long; they only last about a millionth of a
second. It seems absurd and almost impossible that they should be
visible; but in a perfectly darkened room they are visible to an eye
which has been rested in the dark. Upon this thin thread hangs the
success of our undertaking." Multum in parvo, truly!
After a series of preliminary experiments, in which he studied the
various induction effects, including the phenomenon of resonance, and
demonstrated waves on wires (an earlier, but overlooked, discovery of
von Bezold, in 1870), and also solved the problem of the Berlin
Academy— "to establish experimentally any relation between electro-
magnetic forces and the dielectric polarization of insulators"- — he was
fully convinced that the disturbance was propagated through space,
independently of wires, with a finite velocity and in the form of waves,
in accordance with Maxwell's prediction. His conclusion was then
definitely and convincingly proved by making use of the well-known
method of reflection and interference to produce standing waves, and
noting the position of the nodes and antinodes. These epochal experi-
ments were described in a paper entitled "Electromagnetic Waves in
Air and Their Reflection," published in May, 1888. But he did not
stop there, and in these and succeeding investigations he showed that
electric waves are reflected from plane and curved metal surfaces in
accordance with the same laws as light waves; that they are refracted
in passing thorugh prisms of pitch, parafhn, and other dielectrics; and
that they are polarized by a grating of parallel wires, and hence are
transverse waves. From actual measurements of their wave-length
and computations of their frequency (from the constants of his oscil-
lator), he calculated their velocity and found that it was the same as
the velocity of light. As summarized by Hertz himself, "The object
of these experiments was to test the fundamental hypotheses of the
Faraday- Maxwell theory, and the result of the experiments is to con-
firm the fundamental hypotheses of the theory." The old action-at-
a-distance philosophy had come to an end.
The importance of Hertz's contributions to this great subject re-
ceived instant and enthusiastic recognition, and his experiments were
334 BELL SYSTEM TECHNICAL JOURNAL
soon being repeated in all the important laboratories of the world.
The English mathematical physicist, Oliver Heaviside, writing in 1891,
said: "Three years ago electromagnetic waves were nowhere. Shortly
after, they were everywhere." Here were researches of a most abstruse
and complex character, with no apparent utility and having no ele-
ments of popular appeal, and yet bringing to their author such acclaim
as had seldom been accorded to a man of science. Honors were
showered upon him on every hand, at home and abroad. In England,
where his work was especially appreciated, he was awarded the coveted
Rumford medal by the Royal Society.
Hertz's characteristic modesty in referring to his own achievements
was matched only by his generosity in giving credit to the accomplish-
ments of others. In one of his lectures he said, "Such researches as I
have made upon this subject form but a link in a long chain. . . . Lack
of time compels me, against my will, to pass by the researches made by
many other investigators; so that I am not able to show you in how
many ways the path was prepared for my experiments, and how near
several investigators came to performing these experiments them-
selves." Mention has been made of the investigations of Sir Oliver
Lodge in the same field and the imminence of his discovery of the same
phenomena. It is pleasant, indeed, in this instance to be able to record
the absolute lack of any feeling of jealousy or envy on the part of either
of these courteous gentlemen. In the introduction to his collected
papers Hertz wrote, " I may here be permitted to record the good work
done by two English colleagues who at the same time as myself were
striving towards the same end. In the same year in which I carried
jOut the above research. Professor Oliver Lodge, in Liverpool, investi-
gated the theory of the lightning conductor, and in connection with
this carried out a series of experiments on the discharge of small con-
densers which led him on to the observation of oscillations and waves
in wires. Inasmuch as he entirely accepted Maxwell's views and
eagerly strove to verify them, there can scarcely be any doubt that if
I had not anticipated him he would have succeeded in observing waves
in air, and thus also in proving the propagation with time of electric
force. Professor Fitzgerald, in Dublin, had some years before en-
deavored to predict, with the aid of theory, the possibility of such
waves, and to discover the conditions for producing them."
On his part Lodge just as generously wrote, only a few years after-
wards, in an obituary of his rival: "Hertz stepped in before the English
physicists, and brilliantly carried off the prize. He was naturally and
unaffectedly pleased with the reception of his discovery in England,
and his speech on the occasion of the bestowal of the Rumford medal
HERTZ, THE DISCOVERER OF ELECTRIC WAVES 335
by the Royal Society will long be remembered by those who heard it
for its simplehearted enthusiasm and good-feeling. His letters are full
of the same sentiment. ..."
Noteworthy, indeed, was the extreme modesty of this scientific lion
of the hour, and equally striking his consideration for the feelings of
others. It is recorded that when the Royal Society presented him with
the Rumford medal, "he silently disappeared from Bonn for a few days
— none knew why — and he returned as silently." In refusing the
request made by the editor of The Electrician (of London) in 1890 for
his photograph. Hertz replied, " I feel as if presenting my portrait now
in so prominent a place follows too quickly the little work I have done.
I should like to wait a little, and see if the general approbation which
my work meets with is of a lasting kind. Too much honor certainly
does me harm in the eyes of reasonable men, as I have sometimes occa-
sion to observe. If your kind intention is the same in two years, even
one year, I shall readily consent and help you in every respect." Four
years later the portrait was published, following upon Hertz's death.
Upon the untimely ending of his brief but brilliant career, occurring
in the very prime of life, before he was yet thirty-seven, there was a
feeling of shock and sadness in every scientific quarter. Many were
the sincere tributes paid to his memory, honoring him for his rare
personal qualities as well as his distinguished scientific attainments.
Some expressions from Lodge have already been quoted. Said he in
his obituary in The Electrician, "Not a student of physical science on
the planet but will realize and lament the sad loss conveyed by the
message, 'Hertz is dead.'" The editor of that journal wrote, "In the
modesty and self-forgetfulness which blend so admirably in the spirit
of true scientific research Hertz was singularly rich." In an editorial
note in an American journal. The Physical Review, we find the following :
" In addition to the recognition of those who were able to appreciate his
work, Hertz received the acclamations of the entire world of thought.
Fortunately he possessed a nature of such complete simple-mindedness
that his sudden rise into a position akin to notoriety had no effect upon
him. The unassuming bearing which had always characterized him
remained with him to the end."
In a memorial address delivered by Professor Herman Ebert before
the Physical Society of Erlangen on March 7, 1894, the following senti-
ments are expressed: " In him there passed away not only a man of great
learning, but also a noble man, who had the singular good fortune to
find many admirers, but none to hate or envy him; those who came into
personal contact with him were struck by his modesty and charmed by
his amiability. He was a true friend to his friends, a respected teacher
336 BELL SYSTEM TECHNICAL JOURNAL
to his students, who had begun to gather round him in somewhat large
numbers, some of them coming from great distances; and to his family
he was a loving husband and father."
It can be said in retrospect that the fundamental invention in radio-
telegraphy was made by Hertz, and yet it is true that the discoverer
of electric waves had no anticipations as to their utilitarian possi-
bilities. There was no rush to the patent office; indeed, it was not until
two years after Hertz's death that the first application for a radio
patent was filed, by Marconi. The chief interest at the time was purely
scientific, the results being hailed as the settlement of a great scientific
controversy, the confirmation of Maxwell's theory, the annexation to
electricity of the entire domain of light and radiant heat. In the cur-
rent literature we find little of prophecy with respect to utility. Sir
William Crookes has been credited with being one of the first to foresee
distinctly the applicability of "Hertzian" waves to practical teleg-
raphy. In an article in the Fortnightly Review for February, 1892, he
made a remarkably accurate forecast of what was to come: "simpler
and more certain means of generating electrical rays of any desired
wave-length"; "more delicate receivers which will respond to wave-
lengths between certain defined limits and be silent to all others";
"means of darting the sheaf of rays in any desired direction. . . ."
And for secrecy he foresaw that "the rays could be concentrated with
more or less exactness on the receiver," if the sender and receiver were
stationary; or, if moving about, "the correspondents must attune their
instruments to a definite wave-length. . . ." "This is no mere dream
of a visionary philosopher," he wrote. "All the requisites needed to
bring it within the grasp of daily life are well within the possibilities of
discovery, and are so reasonable and so clearly in the path of researches
which are now being actively prosecuted in every capital of Europe that
we may any day expect to hear that they have emerged from the realms
of speculation into those of sober fact."
As we well know, all that he predicted, and more, has become reality,
although progress was not to be as rapid as then seemed probable.
One of those who at the time had the imagination to see, if only hazily
perhaps, the great possibilities of Hertz's discovery was the youthful
Marconi, who had also the initiative and the determination to put his
ideas into execution, to make the new-found waves useful to mankind.
Within a few years, around the turn of the new century, the world was
to be thrilled by the detection of a wireless signal transmitted across
the wide Atlantic. But there were insurmountable limitations to the
means at hand, and it remained for still another wave in the onward roll
of science, the advent of the magical era of electronics, to yield the
HERTZ, THE DISCOVERER OF ELECTRIC WAVES 337
tools necessary for the really great advance that was ahead. With
the invention and development of the amplifying and oscillating
vacuum tube progress was greatly accelerated, and in a comparatively
short time there had been achieved, by a veritable army of experi-
menters, the marvels of world-wide intercommunication which are so
familiar to us today.
Instruments for the New Telephone Sets *
By W. C. JONES
Transmitters and receivers for use at subscribers' telephone
stations have been designed which not only materially improve
transmission but also simplify manufacture and facilitate main-
tenance. This paper discusses these improvements and describes
some of the new design technique employed in their development.
AS a result of continuous development work on transmitters and
receivers for use at subscribers' telephone stations, new instru-
ments have been designed which not only materially improve trans-
mission but also embody features which simplify manufacture and
facilitate maintenance. These instruments are now being produced
for use in handsets, desk stands and wall sets in the Bell System. ^
In many respects these instruments represent outstanding advances
in transmission instrument design and performance. It is the purpose
of this paper to discuss these improvements and to describe some of the
new design technique employed in their development. The data pre-
sented will be confined almost entirely to physical measurements which
serve to define the performance characteristics of the instruments.
The interpretation of these data in terms of their relationship to the
characteristics of associated apparatus and their overall reaction on
transmission in the telephone plant is covered by a companion paper
dealing with the transmission features of the new sets.^
Handset Applications
The new transmitter unit with an adapter was first introduced in
1934 as a replacement for the earlier type of handset transmitter.^
There are now about five million of these transmitters in use in the
plant of the Bell System. While experience has shown that they effect
an outstanding improvement in performance they do not take full
advantage of the possibilities of the unit type of construction from the
standpoint of simplification, owing to the fact that a number of
additional parts are required to mount the unit on the existing type
of handset handle. The advantages of the unit type of instrument
have been realized in a new design of handset introduced during 1937,
about a million of which have been produced. This handset is shown
* Presented at A.I.E.E. Summer Convention, Washington, D. C, June 21, 1938.
338
INSTRUMENTS FOR THE NEW TELEPHONE SETS
339
with the new combined set in the photograph, Fig. 1, and in cross-
section on Fig. 2.
In designing this handset every effort has been made to obtain the
maximum degree of simpHcity consistent with the electrical require-
ments involved and at the same time to secure an attractive design
which harmonizes with the other station apparatus on the subscriber's
premises. Only three phenol plastic parts are employed; namely, the
Fig. 1 — Handset and desk stand equipped with the new instruments.
handle and the transmitter and receiver caps. In designing these
parts particular attention has been paid to providing adequate cross-
sections at the points of maximum stress and to distributing the
weight so as to reduce to a minimum the breaking moments which are
developed when the handset is dropped. The transmitter and re-
ceiver caps serve the dual purpose of holding the units in place and
providing mechanical protection. In addition they thoroughly insu-
late the user from all the metal parts which are included in the electrical
circuit. Both caps have smooth surfaces which can be readily cleaned.
As will be pointed out later, the grid of the receiver cap also has a
transmission function and plays an important part in determining the
response in the upper frequency range. Spring contacts are provided
to facilitate the assembly of the units in the handle. This operation
is further facilitated by the fact that specific alignment of the units
340
BELL SYSTEM TECHNICAL JOURNAL
U
INSTRUMENTS FOR THE NEW TELEPHONE SETS
341
and the caps relative to the handle is unnecessary. The spacing
between the transmitter and receiver is such that the handset can be
used with the existing type of desk mounting as well as with the new
combined set.
Desk Stand and Wall Set Applications
The photograph, Fig. 1, also shows the new transmitter and receiver
unit adapted to desk stand and wall set use. Cross-sections of these
instruments are shown on Fig. 3. The faceplate, mouthpiece and pro-
TRANSMITTER
UNIT
Fig. 3 — Cross-sections of the transmitter and receiver for desk stands and wail sets.
tective grid of the transmitter are combined in one phenol plastic part
which is so designed as to reduce cavity resonance to a minimum and
provide response characteristics essentially the same as those of the
handset transmitter. On the other hand , the mouthpiece is sufficiently
342 BELL SYSTEM TECHNICAL JOURNAL
prominent to encourage the user to talk directly into it and in this
way reduce the losses which often result when flush type faceplates are
employed with desk stand and wall set instruments. A phenol plastic
part, equipped with contact springs, holds the unit tightly in the face-
plate and provides electrical connections.
As in the handset the unit of the receiver is held in place by the cap.
Springs are provided in the shell for bringing out the electrical con-
nections. A metal insert adds sufficient weight to meet the switch-
hook requirements of the existing sets. The phenol plastic parts of
both the receiver and transmitter are so designed as to insulate
thoroughly the units and minimize breakage.
Transmitter Unit
The new transmitter unit is of the "direct action" type, that is, one
in which the movable electrode serves the dual purpose of contact and
pressure surface. As is shown by Fig. 2, this electrode is mounted at
the center of a diaphragm of thin aluminum alloy formed into a shallow
cone and ribbed to add rigidity. "Books" of thin impregnated paper
mounted in a recess in a die-cast frame provide a resilient support for
the edge of the diaphragm. The fixed electrode is held in place in
the frame by a threaded ring and is insulated from the frame by a
phenol fibre washer and a ceramic insulator which also forms one of
the surfaces of the carbon chamber. The active surfaces of both
electrodes are gold plated. A silk annulus clamped at its outer edge
between the ceramic insulator and the frame and its inner edge between
the movable electrode and the diaphragm forms a resilient closure for
the carbon chamber. Electrical connection between the movable
electrode and the frame is provided by means of metal strips of low
stiffness. Provision is made for machine filling the carbon chamber
through a hole in the fixed electrode and closing this hole by means of
a cap which crimps over a projecting shoulder. The exposed surfaces
of the cap and the threaded ring are silver plated and form the contact
surfaces for the electrical connections. A moisture-resistant mem-
brane protects the internal parts of the unit from the effects of con-
densed moisture from the breath. This membrane is clamped at its
outer edge between a protective grid of perforated metal and the frame.
A thin metal ferrule fastens the grid to the frame. The exposed parts
of the unit are anodically finished to resist corrosion.
In addition to being simpler than the earlier transmitter and hence
less difficult to produce, the new transmitter unit has characteristics
such that:
INSTRUMENTS FOR THE NEW TELEPHONE SETS 343
1. Its performance is less affected by angular position.
2. There is less aging under the conditions encountered in service.
3. The electrical output is higher and the response more uniform.
4. The modulation products resulting from non-linearity are materially
reduced.
Effect of Angular Position. — In order to insure good contact between
the carbon granules and the diaphragm in the positions in which the
handset is most likely to be held in service, the carbon chamber of the
earlier transmitter was placed in front instead of the conventional
location in back of the diaphragm.^ The positional characteristics of
this transmitter were further improved by the use of a "barrier" type
of variable resistance element in which the electrodes are stationary
and form the walls of the carbon chamber, and in which the surface of
the diaphragm in contact with the granules is insulated and serves
only as means for changing the contact forces between the granules in
response to the variations in sound pressure at the diaphragm surface.
While this transmitter represented a distinct advance in handset
performance from a transmission standpoint and was quite effective in
reducing undesirable positional effects, particularly in the "horizontal
face-up" position, it was somewhat complicated mechanically and
involved the problem of providing a closure between the diaphragm
and the adjacent electrode which would be sufficiently resilient to
meet the transmission requirements and at the same time prevent
carbon leakage. In addition, there was some degradation in quality
when it was held in the "horizontal face-down" position where the
carbon granules tended to fall away from the diaphragm. While this
condition occurred only infrequently in service, it was one which it was
considered desirable to eliminate if this could be accomplished without
making the structure mechanically complex or difficult to manufacture
or maintain. A tendency also was observed in the field for the re-
sistance to increase sufficiently under certain conditions to react ad-
versely on the operation of the associated signaling apparatus. Owing
to the inherently small areas of the sound passages leading to the
diaphragm the moisture condensed from the breath could not be
excluded by a membrane without complicating the structure and
adding sufficient mechanical impedance to impair transmission.
Following the introductory work on the barrier transmitter, an
intensive study of the direct action type of carbon element was made to
determine whether the limitations of the earlier structures of this type,
which arose from the non-fluid character of the carbon, could be over-
come. This study resulted in the transmitter unit shown on Fig. 2.
This unit eliminates the undesirable features of the inverted type with-
out sacrificing its desirable characteristics.
344
BELL SYSTEM TECHNICAL JOURNAL
The electrode surfaces of the new transmitter unit are so propor-
tioned and so spaced relative to each other that the important current
paths shift their locations in the carbon mass with changes in angular
position in a manner such that the mean effective pressures in the paths
and the lengths of the paths result in substantially constant resistance
in all positions. Furthermore, the components of the axial motion of
the diaphragm effective in changing the contact forces in the paths are
also such as to produce essentially constant modulation. Not only is
the total resistance of the paths between the electrodes substantially
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Fig. 4 — Positional characteristics of the transmitter.
constant, but this resistance also is uniformly distributed between the
individual contacts with the result that at no time does the contact
potential rise to a value sufhciently high to produce objectionable
carbon noise or "burning." These features result in resistance, volume
efficiency and carbon noise characteristics which, as is shown by
Fig. 4, are essentially independent of angular position.
Another and perhaps a more exacting criterion of the adequacy of a
transmitter from the standpoint of its ability to function satisfactorily
in all positions is the extent to which its transmission characteristics
at normal speech intensities are adversely affected when immediately
INSTRUMENTS FOR THE NEW TELEPHONE SETS 345
preceded by loud speech. If a poorly designed transmitter is held in a
position such that the carbon granules tend to fall away from the
movable electrode when this test is applied, the non-fluid action of
the carbon will prevent the reestablishment of contact with the elec-
trode surface with the result that volume losses of as much as 20 db and
a serious degradation in quality take place. Furthermore, these losses
persist until the transmitter is jarred or moved about sufficiently to
change the configuration of the granules. On the other hand, if the
effect of the frictional forces within the granular mass has been taken
fully into account in the design of the carbon element, these forces will
not react in a manner such as to prevent good contact with the electrode
following the large amplitude produced by loud speech and uniform
volume and good quality will obtain at all times. The new transmitter
meets this test with a substantial margin.
Carbon leakage is prevented in the new instrument without impair-
ment of transmission by the resilient silk closure for the carbon
container previously mentioned.
Aging. — Transmitter design has advanced to a stage where heating
at the points of contact in the carbon element need no longer be an
important source of aging. Therefore, such aging of the granular
material as occurs in a well-designed instrument is limited almost
entirely to that resulting from changes in the properties of the granules
caused by abrasion of their surfaces when the transmitter is subjected
to mechanical shocks such as occur when the handset is placed on the
mounting. As in the case of the earlier transmitter, the new trans-
mitter is machine filled ^ with the result that the motion of the granules
and the resultant surface abrasion is reduced to a minimum.
The changes in resistance due to the residual aging have little adverse
effect insofar as volume is concerned. In fact, the constants of most
of the circuits in which the transmitter is used are such that an increase
in resistance adds to rather than decreases the electrical output because
of the greater amount of power supplied to the transmitter from the
central office battery.
On the other hand, an increase in resistance, though small, may
prove to be important in certain circuits where a critical relationship
between transmitter resistance and the performance of associated
apparatus exists. Under these conditions variations in transmitter
resistance may result in failure of the associated apparatus to perform
satisfactorily if certain limiting values of resistance are exceeded. In
determining the limits to be placed on these values account must be
taken of all the variables in the circuit in which the transmitter is
connected. Obviously combinations of variables of this nature
346
BELL SYSTEM TECHNICAL JOURNAL
cannot be dealt with on the basis of averages alone but must include
some measure of their range, such, for example, as the standard
deviation.'* The available data indicate that average transmitter
resistances and standard deviations which lie within the area bounded
by the dotted curve. Fig. 5, will have no adverse effect on circuit
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Fig. 5 — Limiting values of transmitter resistance.
operation in the Bell System plant. This curve is based on certain
marginal circuits of which there are a number in everyday use. An
important transmitter resistance in determining the performance of
associated apparatus in these circuits is the resistance during the
period when the call is being established. This resistance is referred
INSTRUMENTS FOR THE NEW TELEPHONE SETS 347
to as the signaling resistance. As is shown by the soHd curves the
signaling resistance of the earlier type of transmitter after artificial
aging by an amount considered to be the equivalent of four years in
service falls outside the acceptable area. On the other hand, the re-
sistance of the new type when aged and measured under identical
conditions falls well within the limiting curve and hence not only
requires less frequent replacement but also permits greater freedom in
circuit design and plant layout.
Moisture condensed from the breath is an important factor in deter-
mining the life of a transmitter. A protective membrane is provided
in the new transmitter unit which not only is highly moisture resistant
but also results in no appreciable transmission impairment. The
characteristics of the material employed in this membrane are such
that it is not affected by the aging conditions encountered in service
such, for example, as the alkaline reaction of water after it has been in
contact with phenol plastic parts or tobacco ashes. The exposed
metal parts are finished to resist the corrosive action of these agents.
Response. — Reducing the transmitter to an equivalent electrical
circuit provides a useful means for analyzing its performance and deter-
mining the extent to which the individual parts contribute to its
response. Such a circuit for the new unit is shown on Fig. 6.
While the diaphragm can be represented as a lumped mass for
frequencies in the region below 3500 cycles per second, it is necessary
to consider it as being composed of three masses coupled by stiffnesses
in order to represent adequately its performance at higher frequencies.
These masses consist of the central portion m-o, the ribbed intermediate
portion m^ and the outer portion W4. The central portion includes the
mass of the movable electrode and is coupled to the ribbed portion
by the stiffness s^ which in turn is coupled to the outer portion by the
stiffness 52. The paper books which support the edge of the diaphragm
have a stiffness 54 and a resistance ri. Their mass is included in the
mass of the outer portion of the diaphragm W4. The internal re-
sistances of the portions which form the coupling stiffnesses 52 and 56
are represented by ^2 and n respectively. A hole is provided in the
diaphragm to permit rapid equalization of low frequency pressures of
high intensity and prevent damage to the diaphragm and other parts.
The mass and resistance of this hole, nisrs, are so chosen that their
effect on response is confined to frequencies below 300 cycles per second
where the station circuit itself is relatively inefficient. The controlling
stiffness, S3, is that of the cavity between the diaphragm and the die-
cast frame. As is to be expected the impedance of the carbon granules
is a function of amplitude and frequency. However, for the purpose
348
BELL SYSTEM TECHNICAL JOURNAL
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INSTRUMENTS FOR THE NEW TELEPHONE SETS 353
3. These improvements have been accomplished without sacrificing
simpHcity of design or introducing features which complicate
manufacture of the receiver or increase the maintenance
required.
Response. — An equivalent electrical circuit for the receiver and a
typical closed coupler response curve are shown on Fig. 8. Referring
to this figure it will be noted that there are two meshes in the circuit
which contain mass, stiffness and resistance and which control the
motion of the diaphragm. One of these meshes consists of the acous-
tical resistance, Wi^i, coupled to the diaphragm, maSafo, by the stiffness,
5i, of the cavity between the diaphragm and the plate which surrounds
the pole tips. Included in this mesh is the stiffness, s^, of the cavity
in the handset handle or receiver shell. The other mesh is composed
of a cap grid, mzYz, and the load, Si, coupled to the diaphragm by means
of the cavity stiffness, S3. The grid of the receiver unit proper is
provided for mechanical protection only and has holes large enough to
have no reaction on response. The mass of the resilient screen is
small and is lumped with the diaphragm mass, mo. The electrical
portion of the circuit consisting of the winding, RiLi, and the equiva-
lent eddy current circuit, R^Li, is coupled to the mechanical and
acoustical portion by means of the force factor M''.
The response computed from the equivalent circuit for a number of
frequencies is included on Fig. 8. The agreement between this curve
and the measured curve is excellent and makes it possible to predeter-
mine the response of the receiver with a high degree of accuracy, and to
evaluate the effect on the overall response of the receiver of changes in
the constants of the component parts. This type of analysis also has
been invaluable as an aid in establishing the causes of variations in
response which have been observed during the development and pro-
duction of the receiver. A measured response curve of a receiver of
the earlier type has been added to Fig. 8 for convenience of reference.
The improvement in uniformity and range of response is obvious. It
will be noted that large gains have been effected for frequencies in the
range from 1500 to 3000 cycles per second.
The response of the receiver to a square topped wave affords an excel-
lent measure of frequency distortion. Oscillographic records of the out-
put of typical receivers of the new and earlier types are shown on Fig. 9
for a frequency of approximately 50 cycles per second. The distorting
effect of diaphragm resonance is so obvious as to require no comment
beyond pointing out that for accurate reproduction of square waves
uniform response for an infinite frequency range is required and that
the slight rounding of the corners of the wave as reproduced by the
354
BELL SYSTEM TECHNICAL JOURNAL
aoiSJCOND |.^;^
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Fig. 9 — Response of the receiver to square topped waves.
A — Measuring circuit — no receiver.
B — Early type receiver.
C — New type receiver.
new receiver is due primarily to the falling off of its response above
3000 cycles per second.
The substantially uniform response of the new receiver also renders
clicks and other surges much less objectionable. This is due to the
fact that the ear does not respond to the peak value of an oscillatory
transient alone but integrates the oscillation over an interval at the
beginning of the surge, hence the higher the damping the less ob-
jectionable the click.
The non-linear distortion produced by a receiver of the new type is
negligible in its reaction on transmission, the harmonics in the output
being 35 db or more below the fundamental.
Magnetic Circuit. — Inasmuch as the magnetic properties of the
diaphragm, as well as its mechanical properties, must be considered in
arriving at the preferred dimensions, it was necessary in designing the
new receiver to develop criteria which could be applied in determining
the optimum relationships between these factors. This study led to
the use of the ratio of the force factor to the effective mass of the
diaphragm for this purpose. For given magnetic materials in the pole-
pieces and the diaphragm and a given air-gap length, there is a pole
face area and diaphragm thickness for which this ratio is a maximum.
Typical data illustrating this relationship are shown on Fig. 10. The
available magnetic materials were studied using this technique and a
decision was reached to use permendur in the diaphragm and 45 per
cent permalloy in the pole-pieces.
INSTRUMENTS FOR THE NEW TELEPHONE SETS
355
There is a value of polarizing flux for which the force factor of the
given magnetic circuit is a maximum. The rate at which the force
factor falls off above and below this optimum value of flux is a function
of the magnetic characteristics of the materials employed, the length
Fig. 10 — Force-to-mass ratio as a function of diaphragm thickness and pole-piece area.
of the air-gaps, etc. Without exception the more efficient magnetic
circuits have been found to be the most critical as regards polarizing
flux. Hence, if wide variations in the efficiency of the product
receivers are to be avoided and serious losses due to subsequent demag-
netization in service prevented, means must be provided not only for
bringing the flux in each receiver to the optimum value, but also for
insuring that it remain at this value during the life of the instrument.
In order to accomplish this result the magnets of the new receiver are
so designed as to overpolarize the magnetic circuit when they are fully
magnetized. Equipment is provided for demagnetizing each receiver
to its optimum flux value during the assembly process. Receivers
356 BELL SYSTEM TECHNICAL JOURNAL
which are not sufficiently overpolarized before demagnetization to
resist further demagnetization under service conditions are rejected.
Temperature Effects. — The diaphragm of the new receiver is held in
place by the force developed by the polarizing flux and hence it is free
to expand and contract independently of its seating surface. This
feature renders the performance of the receiver independent of the
changes in temperature to which it has been subjected. The force due
to the polarizing flux is sufficiently high to prevent rattling at input
intensities many times those of loud speech.
Coupling
Although station circuits can be designed which under ideal con-
ditions result in no sidetone, this objective is never fully realized under
actual plant conditions, with the result that a part of the electrical
output from the transmitter always reaches the local receiver.
Whether the electrical coupling between the transmitter and receiver
as evidenced by the residual sidetone is of importance from the stand-
point of sustained oscillation or "howling," depends upon the degree
of mechanical and acoustical coupling between the instruments.
Handset and instrument design has advanced to a stage where mechan-
ical coupling need no longer be a problem. On the other hand, as
the response of the instruments is improved, the acoustical coupling
may become an important item in determining the howling margin.
This margin is so large under the conditions where the new handset is
being used for transmission purposes that there is no tendency for
oscillation or distortion to occur. However, if the handset is placed
face downward on a desk or table, an air column is created which
resonates in the region of 2500 cycles per second. Inasmuch as this
is the region where a substantial improvement in the response of the
receiver has been efifected, a marked reduction in howling margin
results. While there is still sufficient margin to meet all of the require-
ments of field use, this situation serves to emphasize the fact that such
factors as acoustic coupling may limit the transmission improvements
which can be efifected under a given set of operating conditions.
Effective Transmission
The extent to which the better performance of the new instruments
is effective in improving the grade of transmission afforded the tele-
phone user is a complex matter and one which is influenced by such
factors as the characteristics of the circuits with which the instruments
are associated at a given time, the amount of noise present at the
transmitting and receiving stations, the reaction of sidetone on the
INSTRUMENTS FOR THE NEW TELEPHONE SETS 357
loudness with which the user speaks, the distance between his Hps and
the face of the transmitter, the tightness with which he holds the
receiver to his ear, etc. Many of these factors are beyond the control
of the engineer responsible for the design of the transmitter and
receiver and hence can be evaluated, insofar as their reaction on trans-
mission is concerned, only by tests made under the conditions of
actual use.
A method has been devised which makes it possible to rate the over-
all effect of these factors on transmission in a way representative of the
results obtained by the subscribers in their normal use of the instru-
ments.^ Numerous tests employing this method of rating were made
during the development of the new transmitter and receiver to make
certain that the course followed in their development would insure the
best possible performance under service conditions. Similar tests
were also made of the designs selected for production. These tests
show that in many respects the new instruments represent outstanding
advances in transmission instrument design and performance.
References
1. "Scientific Research Applied to the Telephone Transmitter and Receiver," E. H.
Colpitts, Bell System Technical Journal, volume 16, July 1937, pages 251-274.
2. "Transmission Features of the New Telephone Sets," A. H. Inglis, this issue of
the Bell System Technical Journal.
3. "Development of a Handset for Telephone Stations," W. C. Jones and A. H.
Inglis, Bell System Technical Journal, volume 11, April 1932, pages 245-263.
4. "An Introduction to the Theory of Statistics," G. U. Yule, J. B. Lippincott
Company, 1916, page 134.
5. "A Voice and Ear for Telephone Measurements," A. H. Inglis, C. H. G. Gray and
R. T. Jenkins, Bell System Technical Journal, volume 11, April 1932, pages
'293-317.
6. "Magnetic Alloys of Iron, Nickel and Cobalt in Communication Circuits," G. W.
Elmen, Electrical Engineering, volume 54, December 1935, pages 1292-1299.
7. "Electrical Vibration Instruments," A. E. Kennelly, The Macmillan Company,
1923, page 88.
8. "Rating the Transmission Performance of Telephone Circuits," W. H. Martin,
Bell System Technical Journal, volume 10, January 1931, pages 116-131.
Transmission Features of the New Telephone Sets *
By A. H. INGLIS
The new telephone instruments now being introduced by the
Bell System result in an outstanding improvement in transmission
performance in service. The evidence for this, as obtained by
comprehensive laboratory and field tests, is presented here together
with a discussion of the factors responsible for this superior per-
formance and of the consideration involved in its appraisal.
NEW telephone instruments are being applied in the plant of the
Bell System to the deskstand, wallset and handset, and result in
markedly improved transmission performance. The new instruments
are associated with the anti-sidetone feature which is also applied to
the older sets already in plant. The selection of these particular
designs from the wide choice made possible by new design technique,
materials and manufacturing methods, has been based on develop-
ments in the methods for quantitatively rating the relative merits of
different designs. In general there has been consistent effort over a
period of years to base these ratings primarily on performance in
service rather than on laboratory tests.
The factors influencing service performance are so many, and so
complicated in their relationship, and are in so many cases difficult or
even impossible for the designer to evaluate or control, that their net
effect on performance cannot be predicted with certainty by laboratory
methods. Of necessity such methods involve a limited selection of
primary test conditions, and an even more limited selection from the
possible combination of these conditions. This Is particularly true
in the rating of the transmission performance of a telephone set.
Laboratory tests are essential in the study and analysis of design
problems, and are invaluable similarly in interpolating, supplementing,
and explaining service performance results. In determining the reac-
tion on the user of the transmission features of possible designs,
however, the field performance test has been found of first Importance
in deciding what particular characteristics to include in the new tele-
phone instruments and circuits.
* Presented at A. I. E. E. Summer Convention, Washington, D. C, June 21, 1938
358
TRANSMISSION FEATURES OF NEW TELEPHONE SETS 359
Important Transmission Characteristics
OF THE New Telephone Sets
The specific transmission design features of the new instruments are
described elsewhere.^ The purpose here, therefore, is to discuss pri-
marily the outstanding improvements in performance resulting from
the application of the new instruments and the anti-sidetone feature
which has been available for some time.
These improvements are
1. Those due to the station circuit, which, as compared with the
previous station circuit,
a — largely reduce the efficiency of the sidetone path between
transmitter and receiver without materially affecting the
electrical efficiency of the set in transmitting or receiving.
This means that sounds, either noise or speech, which are
picked up by the transmitter are reproduced in the receiver
of the same set at a much lower level.
b — reduce the susceptiveness for certain types of party line
sets to interference with reception by noise set up by power
transmission systems.
2. Those due to the physical characteristics of the transmitter and
receiver.
Fig. 1 — The new handset and deskstand telephone instruments.
Several of these features have been available for some time and
have, of course, been introduced into the plant as they became avail-
able. The new transmitter and the anti-sidetone circuit, for example,
have been standard for some years and have already been installed in
large numbers.
360
BELL SYSTEM TECHNICAL JOURNAL
Figure 1 shows both the new handset and the deskstand forms of
mounting, including all these features as integral parts of their design.
The new desk type transmitter and receiver can, of course, be used
with wall sets.
The schematic drawing of Fig. 2 indicates the general arrangement
of parts in the new station transmission circuit for either type of set.
(So —
ANTI-SIDETONE
INDUCTION COIL
— Ksmj — '
^
■SWITCH-HOOK'^
CONTACTS
xs
Fig. 2— Schematic transmission circuit of anti-sicletone coil.
In describing the results produced by these transmission features,
and the methods employed in measuring and rating these results, it
seems desirable to include some discussion of the characteristics of a
telephone conversation as distinguished from a direct, face-to-face
conversation, so that the various effects of the new circuits and
instruments may be seen in as correct relative proportion and as
generally comprehensible form as possible.
Some Elements of the Station Transmission Problem
In either a telephone or a direct conversation, successful cummuni-
cation depends on the characteristics of the talker and of the listener,
and their reactions to each other and to the character of their sur-
roundings. In a direct conversation such, for example, as across a
desk, the environment is in general the same for both talker and
listener, and their ears are materially aided by their eyes. In a
telephone conversation, however, not only may the surroundings of
talker and listener be entirely different, but a third element, the
telephone system, is added to the environment of each user, which
complicates his reaction, not only to his own surroundings, but also
to the other party to the conversation. Furthermore, for obvious
economic reasons, the natural binaural reception of direct conversa-
tion, with its advantages in discriminating between sounds from dif-
ferent directions, is replaced in the telephone conversation by a
monaural medium.
TRANSMISSION FEATURES OF NEW TELEPHONE SETS 361
Fundamental differences of this kind between telephone and direct
conversation must be taken into account in the design of a telephone
transmission system if satisfactory results are to be obtained. For
example, the talker is accustomed in a direct conversation to regulate
his talking volume by what he himself hears under prevailing noise
conditions (which incidentally are the same for the listener), by the
ease with which he hears the other party, and by the ease with which
the listener appears to hear him. By experience, under ordinary
conditions, the first factor mentioned, the loudness with which the
talker hears himself, probably comes to be the primary control on his
talking volume.
These various factors also serve to regulate talking volumes in
conversation by telephone, but their magnitudes and the relations
between them differ from the condition of face-to-face air path con-
versation and vary from one type of telephone connection to another.
For example, the "sidetone" of the telephone set, being materially
higher than the air-path sidetone, deceives the talker, not only by
making him think he is talking louder than he really is, but also by
apparently modifying the noise conditions under which he is talking
in the pickup and amplification of room noise by his telephone trans-
mitter. Since, in addition the efficiency of the telephone circuit itself
may be different in the two directions of transmission, the loudness
heard by one party may differ more from that heard by the other than
in the case of air transmission. Then, too, noise conditions may be
and frequently are quite different at the two ends of the telephone
circuit. Figure 3A shows the probability of noise of various average
intensities at subscribers' stations as determined by several surveys
covering a large number of locations. On the assumption that any
one of the stations represented by these data may with equal prob-
ability call any other one. Fig. SB has been computed, showing the
probability of noise at the two stations of a telephone connection
differing by more than a certain amount. It will be noted that there
is about an even chance of the noise at the two ends differing by more
than 12 db. In view of these differences, a person's judgment of how
well he is heard and understood can not be as direct as in the case of
air transmission.
In addition, the transmission over the commercial telephone system
affects the quality of the received speech more than the usual room
surroundings in air-path transmission. While acoustic resonance and
reverberation in a room do distort speech, in the extreme case to a
point where understanding may be difficult, such a condition is dis-
tinctly unusual. Equal freedom from distortion in a telephone system
362
BELL SYSTEM TECHNICAL JOURNAL
is a more difficult and expensive condition to obtain than in direct
conversation a few feet from a listener. Something less than perfect
reproduction must suffice, for the present anyway, if costs are not to
be prohibitive.
90
80
70
60
50
40
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sound level in decibels
(above io"'6 watts per cm2)
35
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difference in decibels (between
room noise at ends of telephone
connection)
Fig. 3 — Noise conditions at telephone stations.
All of these differences involve the acquiring by the user of a set
of telephone habits which differ from' those he has acquired in direct
conversation. The problem of the transmission design of a practical
telephone system requires, then, for a satisfactory solution, not only
a determination of the proper speech levels to be delivered, and of the
sidetone characteristics which will, under the conditions of a telephone
conversation, give optimum results with the noise encountered, but
also a decision as to what particular frequency range and characteristic
to choose. Properly designed, a telephone transmission system should
minimize, to the degree consistent with costs, its inherent differences
from direct conversation, and make it easy for the ordinary user to
get, without undue effort, results which are satisfactory to him in
comparison with direct conversation.
In the earlier days of telephony, the problem presented appeared
much simpler. It was, in effect, uni-dimensional, calling primarily for
more efficient instruments and circuits ; more and more power delivered
to the listener's ear. While methods for the control of sidetone were
TRANSMISSION FEATURES OF NEW TELEPHONE SETS 363
not unknown, the importance of such control was not fully appreciated.
Little choice was available in the quality of reproduction provided by
transmitter and receiver, because of meager design knowledge.
Relatively recent and quite rapid developments in knowledge of the
problems involved, in materials, methods, and measuring facilities,
have now presented the necessity for a solution in essentially a three-
dimensional form. These three dimensions may be described as
volume, noise and quality. The solution of the problem on this
basis is obviously more difficult, and has required the development
of methods for quantitatively evaluating and rating their character-
istics in terms of some common yardstick.
Methods of Rating Transmission
For reasons already suggested, such a yardstick must be based on
service performance — on the results obtained by actual users in the
course of day-to-day telephone service.
Extensive investigation has indicated that the best comparative
measure of this transmission performance in local exchange service is
to be found in the time rate of the occurrence of repetitions required
by subscribers for understanding telephone conversations.^ Or, more
explicitly, when two transmission conditions have the same repetition
rates, all other service factors being equal, these conditions are taken
to be equal with respect to transmission performance. Where two
conditions are not alike it is usually possible to evaluate the difference
in the repetition rates for the same users by inserting distortionless
loss in the better condition until both have equal repetition rates.
Thus, by taking as a reference a typical telephone circuit of specified
make-up, the effects of various factors such as distortion, noise,
attenuation, sidetone, or type of instrument, may all be expressed in
the common terms of the reference circuit trunk which will give the
same repetition rate.
Instead of making this adjustment in every case for the purpose
of evaluating the relative performance of different test conditions for
the same users, the evaluation may be made rather closely over a
limited range by the following typical relation derived from repetition
observations on circuits containing trunks, the losses of which were
varied over a range of values.
db = 50 logio RilR2,^
where i?i, and R2 are the repetition rates of two conditions under com-
parison, and the db figure is the change in the reference trunk which
has the same relative effect on the repetition rate.
364 BELL SYSTEM TECHNICAL JOURNAL
Such a method is, of course, somewhat cumbersome, and requires
a large amount of data to iron out random variations and individual
pecuHarities of Httle general interest. But as the fundamental rating
method, supplemented by laboratory test, it has been systematically
used in studying the value of the anti-sidetone circuit and in selecting
instrument characteristics.
Supplementing the repetition observations, it has been found useful
in service rating to obtain data on speech levels delivered to the line
for each condition observed. This has been done with the volume
indicator, a vacuum tube voltmeter so designed that the reading is
approximately proportional to mean syllabic voltage.'* The informa-
tion thus obtained is useful not only in analyzing the results of service
tests but also in determining typical values for speech levels, necessary
for laboratory tests.
Laboratory tests are of two general types: objective measurements,
and subjective tests. Transmission measurements cover a wide field
with objectives ranging from the physical analysis and study of dif-
ferent designs, to the determination of overall performance character-
istics of structures and systems. It is these latter tests that we are
more particularly interested in here, as most descriptive of the physical
properties of importance in providing telephone transmission service.
Subjective tests in the laboratory may be said to be midway between
physical measurements and field performance tests. Made under
controlled and somewhat artificial conditions, they indicate quanti-
tatively the capabilities of a telephone system in transmitting articulate
speech under the particular conditions of the test. They cannot, of
course, indicate the relative probability of occurrence, and hence
importance, of these different conditions, nor predetermine how well
the subscriber will avail himself of the capabilities provided.
Consideration of some of the results of investigations in both labora-
tory and field will do much to explain the rather large transmission
improvement realized by the introduction of the new sets in actual
service, particularly if examined with the conditions of a direct con-
versation as a basis of comparison.
The Station Circuit
There are two characteristics of the new station circuit of particular
importance from a transmission standpoint.
Reduction of Sidetone
The first is the anti-sidetone induction coil through which the
transmitter and receiver are coupled to the line. This coil comprises.
TRANSMISSION FEATURES OF NEW TELEPHONE SETS 365
in addition to three transformer windings, a balancing network. The
circuit, made up of the four elements: transmitter, receiver, line, and
network, coupled by the transformer, functions in such a manner that
the transmitter and receiver are in conjugate relationship, i.e., voltages
produced by the transmitter are balanced out and do not affect the
receiver. Theoretically, such a circuit, with pure resistance elements,
can be perfectly balanced at all frequencies with complete elimination
of sidetone, and at the same time be as efificient as can any transformer
coupling in an invariable telephone set,* for the transfer of power from
the transmitter to the line, and from the line to the receiver.
This type of circuit is not new in principle, and many varieties are
known and have been described.^ Many of these arrangements, for
one reason or another, are not suitable for application. Some, for
example, call for impedances of transmitters or receivers differing
widely from those available. Certain others are not economical for
common battery service, where the transmitter must receive its battery
supply from the line. Still others require relatively complicated and
expensive cording and switchhook arrangements. The circuit which
has been chosen for general common battery subscriber station applica-
tion, and shown schematically in Fig. 2, is not only as simple and as
easily adapted to Bell System conditions as any, but permits a coil
design which is economical to manufacture as well as efificient in
performance. Other types of anti-sidetone circuit have been adopted
for local battery station service and for operators' telephone sets.
The theory of operation of this anti-sidetone circuit has already
been discussed elsewhere.^ It is intended here to show the general
purposes of the application, some of the considerations involved in the
design, and the kind of results accomplished.
While in theory complete elimination of sidetone is possible, as well
as ideal efficiency of transformation, in practice neither objective
can be entirely realized. The unavoidably wide variations in line
impedance looking from the set, ranging from high positive to high
negative phase angle, and from a few hundred to more than a thousand
ohms in magnitude, together with other practical departures from
ideal conditions, necessitate a choice between a high degree of side-
tone balance and the standardization of a minimum number of coil
designs. The variations in loop length and resistance, by their effect
on transmitter battery supply, and consequently on transmitter re-
sistance, furthermore cause variations in the absolute transmitting
and sidetone efficiency of the terminal set, which must be taken into
account in the station circuit design.
The actual design chosen is so arranged as to favor sidetone balance
366 BELL SYSTEM TECHNICAL JOURNAL
on average and shorter loop conditions where transmitter battery
supply is greater, with consequent higher sidetone, and to favor trans-
mitting and receiving efficiency on longer loops where battery supply
is low. Since loop losses are greater for transmitting than for receiving
because of transmitter battery supply loss, the ratio of the transformer
is such as to favor the transmitting efficiency of the set somewhat in
comparison to the receiving efficiency. This has the advantage of
raising the transmitted speech level further above line noise. The
same idea, of course, was followed in the design of the sidetone set.
The resultant anti-sidetone circuit adopted and here discussed, as
compared with the sidetone circuit previously in general use, when
equipped with the same transmitter and receiver and on the same loop
and trunk, reduces sidetone on the average by about 10 db. Under
the most unfavorable conditions of use, the reduction is unlikely to
be less than about 7 db compared with the corresponding sidetone
connection. Under the best conditions of balance encountered the
reduction may be as much as 12 db. On the effective basis of trans-
mission the average net improvement in transmission which results is
about 6 db.
From the electrical circuit standpoint alone, the efficiency of the
anti-sidetone arrangement is below that of the sidetone set in the
order of about one or two db in transmitting and in receiving, which
is necessitated by the limitations of practical design and circuit con-
ditions discussed above.
Figures 4a and 46 show for transmitting and receiving, respectively,
the difference in efficiency, with respect to frequency, of the anti-
sidetone set from the sidetone set, each with the same instruments.
Two subscriber loop and trunk conditions are shown : an average loop
and trunk, and a long cable connection.
Figure 4c shows the variation in sidetone reduction with frequency,
of the new set as compared with the corresponding standard sidetone
set, for the same two circuit conditions as above. The curves are
indicative of the effect of variation of circuit impedance on sidetone
balance, in changing not only the magnitude, but the frequency range
in which the best balance occurs.
Data of this sort alone do not, of course, indicate the relative
transmission performance of the two sets. The beneficial effect on
the telephone user of the large reduction in sidetone must be evaluated
on the same yardstick as the losses in transmitting and receiving
efficiencies which, in the practical case, accompany this reduction in
sidetone. McKown and Emling have shown the effect of changes of
this sort on the results obtained by the ordinary telephone user, in
TRANSMISSION FEATURES OF NEW TELEPHONE SETS 367
terms of net effective transmitting and receiving loss, as determined
by service observations.* Their data, shown in Fig. 5, are relative
to the sidetone of a reference set. The heavy solid lines are the
original experimental data, the dotted extensions to these curves
being extrapolated.
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f \
A — TRANSMITTING
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B — RECEIVING
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1-MILE 24-GAUGE LOOPS, 900-OHM TRUNK
5-MILE 19-GAUGE LOOPS, 6-MILE 22-GAUGE TRUNK
SAME SETS AT BOTH ENDS
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—
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-20
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0.5
1.0 1.5 2.0 2.5 3.0
FREQUENCY IN KILOCYCLES PER SECOND
4.0
Fig. 4 — Circuit efficiency of anti-sidetone circuit vs. sidetone circuit.
In addition to the original ordinates, others are shown which are
of interest. They are based on the results of loudness balance tests,
and while not perhaps of great precision, do approximately indicate
the relationship of the sidetone of a telephone conversation to that
for direct speech, and illustrate the differences in the effects of sidetone
for transmitting and for receiving.
On each curve are indicated the average sidetone value of the
standard sidetone and the new anti-sidetone set, each, as before, with
the new transmitter and receiver. There is also shown the range of
sidetone for each type of set, within which practically all service
conditions will fall. This indicated range takes into account not
only variations in sidetone balance due to line impedance variations,
but changes (with loop resistance) in battery supply to the trans-
mitter. It should be noted that in only a few cases is the absolute
sidetone of the anti-sidetone set on the worst sidetone conditions, as
high as or higher than that of the sidetone set on the best sidetone
368
BELL SYSTEM TECHNICAL [JOURNAL
conditions, and then only by a small amount. Furthermore, in spite
of the wider variations in sidetone of the anti-sidetone set, these varia-
tions are over a range such that the resultant variations in effective
losses are smaller than for the sidetone set.
Considering Fig. 5a, it will be noted that for either sidetone or
anti-sidetone sets, the sidetone is louder than for natural speech
RANGE OF
SIDETONE SET u-
I
AVERAGE I
35 30 I 25
TELEPHONE-SET SPEECH-SIDETONE
20
IN DEC
I
_ RANGE OF ,
ANTI-SIDETONE SET
AVERAGE
15 I 10 I 5 0
IBELS (above air SPEECH-SIDETONE)
15 10 5 0 -5 -10 -15 -20
TELEPHONE-SET SIDETONE ROOM-NOISE IN DECIBELS
(above ROOM-NOISE IN FREE EAR)
Fig. 5 — Effects of sidetone on user of telephone.
sidetone, which, as noted before makes the user think he is talking
louder than he actually is. The average sidetone reduction of 10 db
for the anti-sidetone set results in less of this restraint on his talking
level, with a resultant net effective gain in transmitting of about 4
db compared with the sidetone set.
In receiving, Fig. Sh, sidetone introduces an effective loss by the
reproduction in the telephone ear of room noise picked up by the
transmitter. It will be noted that for the anti-sidetone set, the
reproduced noise is in general appreciably lower than the room noise
TRANSMISSION FEATURES OF NEW TELEPHONE SETS 369
itself. Inasmuch as room noise also interferes directly with received
speech in the telephone ear by leakage under the receiver cap, the
contribution to the total noise of the sidetone pickup of the anti-
sidetone set is in most cases small. This is not true for the sidetone
set, where in many cases the sidetone noise may constitute the prin-
cipal interfering noise. The resultant net effective gain in receiving
is about 2 db compared with the sidetone set.
This is a good illustration of the type of information which can be
obtained only from a field study. For example, the relationships
indicated on Fig. 5 are dependent on how far away from the mouth-
piece of the transmitter and at what level the speaker talks, and on
how tightly to his ear he holds the receiver. These in turn are result-
ants of all the conditions of the particular telephone conversation.
If incoming levels are so high as to be uncomfortable, the receiver may
well be held farther away from the ear. In that event, of course, the
sidetone conditions of the set become relatively less controlling. The
weight, size, and shape of the instrument in his hands may similarly
affect the subscriber's use of it, the results he gets, and the relative
importance of various factors of telephone design.
For such reasons, not only must laboratory performance tests be
supplementary and subsidiary to field tests, but additional field tests
must be the basis for determining the effect of any major changes in
design, whether or not those changes are electrical, acoustical, or
purely mechanical.
Considerations of this sort emphasize the importance of having
clearly and explicitly in mind the conditions and relationships of
direct conversation, as a general reference for the interpretation and
explanation of the effects of telephone design on telephone conversa-
tions. The sidetone ordinates of Fig. 5, for example, not only suggest
the difference in function of the anti-sidetone circuit in transmitting
and receiving, but also emphasize the fact that the overall sidetone
resulting from the combination of circuit, instruments, and method of
use, is the important factor rather than the sidetone circuit efficiency
only. Such matters are easily lost sight of, if design is not properly
coordinated in its correct perspective.
The reduction of sidetone provided by the anti-sidetone sets is of
further advantage in two rather different ways.
In attaching a transmitter (which is an amplifier) and a receiver,
to a common handle which mechanically couples the two, a condition
is set up in which the gain under certain conditions may exceed the
loss in the path made up of handle, air, and electrical sidetone circuit.
Sustained oscillation, or howling, will then result between transmitter
370 BELL SYSTEM TECHNICAL JOURNAL
and receiver. Even if this point is not reached, but is approached
within 6 db or so, impairment of quahty results from incipient oscilla-
tion. The greater sidetone circuit loss of the anti-sidetone circuit
provides an additional margin of safety against any such condition.
The granular carbon of the transmitter, and the design of the
transmitter itself must be carefully controlled, or serious noise —
transmitter "burning" — will cause noise in the receiver of the set.
The mechanical and electrical wear and tear of service tend to make
this transmitter noise worse. In the new transmitters this "burning"
has been kept at a low inherent value throughout life. The anti-
sidetone circuit, however, provides a margin of safety against the
small amount remaining, so that with this set there is less likelihood
of transmitter noise causing impairment of reception.
Reduced Susceptiveness to Interference
It will be noted from the schematic circuit drawing Fig. 2 that two
condensers are used in the new sets, one in the anti-sidetone trans-
mission circuit, and a separate one with the ringer. In some types
of party line practice the ringer of the set is connected for some
parties from one side of the line, and for the others from the other
side to ground.
Figure 6 shows schematically two such ringing arrangements during
the conditions of conversation, 6a as used in the new sets, and 66
with one condenser common to transmission and ringing circuits.
It will be noted that in the standard circuit adopted (Fig. 6a), if any
longitudinal noise voltages exist between the central office and station
grounds, there is an equal voltage drop from each side of the line to
ground through the ringing paths (assuming the two ringer condenser
paths to be identical). The voltage drop across the terminals of the
talking set is therefore zero and no noise results.
If the arrangement of Fig. dh, corresponding closely to previous
designs, were used, however, this condition would not obtain. The
condenser of the station in use being common to the transmission
circuit as well as the ringing circuit, the noise voltage drop across this
condenser is introduced in the transmission circuit. In addition there
are other paths to ground from each side of the line through the
transmission circuit which are not of equal impedance. The net result
is a residual noise current through the receiver of the talking circuit.
In the actual case, the impedance of all ringers and condensers is
not identical and there are often more parties connected to one side
of the line than the other. Even under these relatively unfavorable
conditions, however, the two-condenser arrangement adopted reduces
TRANSMISSION FEATURES OF NEW TELEPHONE SETS 371
the susceptiveness of the set to interfering noise by as much as 15 db.
A further material reduction is reaHzed by the high impedance of the
ringer used in the new sets, so that in most cases interfering noise at
grounded ringer stations will not dififer materially from that at indi-
vidual stations where the ringer is bridged across the line.
STATION IN USE
I CORD
LOOP CIRCUIT
B - POSSIBLE SINGLE-CONDENSER ARRANGEMENT
Fig. 6 — Ringing arrangements for party line service.
It is interesting to note that this improvement is realized at little
additional cost, since the transmission condenser, which must be of
relatively high capacitance, is permanently bridged by the transmitter,
so that it is protected from exposure to any large voltages, and may
be of cheaper construction and smaller in size than would otherwise
be the case. The ringing condenser on the other hand, while it must
be constructed to withstand higher voltages, may be of relatively
small capacitance, which gives more uniform and better ringing and
dialing performance.
Characteristics of Transmitter and Receiver
Since the individual design characteristics of the new transmitter
and receiver are discussed elsewhere,^ attention here will be centered
on the overall effects of these characteristics in the complete trans-
372 BELL SYSTEM TECHNICAL JOURNAL
mission system, as indicated both by laboratory and by field test.
As stated before, the problem may be more or less arbitrarily separated
into three correlated problems — volume, quality and noise.
As in the case of sidetone, these problems appear, perhaps, more
nearly in a proper perspective if considered in comparison with the
corresponding factors in a direct conversation. It must be remem-
bered that telephone service does not consist in the provision of a
mechanism, per se, but in the provision of facilities for conversation,
to which the mechanism should be incidental, however important.
Since the inherent conditions of such a conversation are quite different
in many respects from those of a direct conversation with which,
consciously or unconsciously, it will be compared in its overall results,
the parallelism in detail should not be too exact. Departures from
the conditions of direct conversation in certain respects which are
relatively unavoidable, may be best compensated for by deliberate
departure in certain other respects. For example, the physical absence
of one party to the telephone conversation, and the monaural nature
of such a conversation, may be partially compensated for by delivering
to the ear of the listener a somewhat higher speech level than he is
accustomed to in direct conversation. The limitation of frequency
band width imposed on the telephone medium, largely for economic
reasons, may be minimized in its effects if the transmission character-
istics in the available band are other than a facsimile of the corre-
sponding band in direct conversation. All such measures must be
employed with knowledge of their effect on the ultimate objective,
that the telephone conversation may be easy and natural.
General Requirements ^
It is easily seen that for any particular overall frequency char-
acteristic of a telephone transmission system, there are practically an
infinite number of ways in which it can be split up between trans-
mitting and receiving characteristics. From this standpoint alone,
then, there is no particular "best" transmitter or receiver frequency
response. From other standpoints, however, certain general types of
individual characteristics, both in frequency and efficiency, are to be
preferred to others, particularly when considered in their practical
application to an already existing telephone system. It has been
pointed out ^ that in general, development has been toward a telephone
system where both transmitter and receiver are relatively uniform in
their frequency characteristics. Induced noise appears to be so evenly
distributed with frequency that such response would not appear to
magnify the interference problem.
TRANSMISSION FEATURES OF NEW TELEPHONE SETS 373
Transmitting efficiency should be as high as required to keep the
speech well above induced noise but not so high as to cause excessive
crosstalk into other telephone circuits. The maximum desirable re-
ceived level is determined for a given telephone system by the limita-
tions of the human ear in accepting with comfort speech levels above
a certain intensity. Finally, the practical necessity of working as
satisfactorily as possible in conjunction with the telephone trans-
mitters, receivers, and sets in the existing plant during the period of
transition, places a practical limitation on the amount of change that
is desirable in relative levels of either transmitting or receiving.
With regard to frequency range, previous work ^° indicated the
desirability of designing circuits to transmit frequencies from 200 or
300 cycles up to about 3,000 cycles. Gains in articulation and natural-
ness are realized by increases in this band width, but are progressively
smaller for successive equal increments in frequency. A 3,000-cycle
band properly used gives good transmission both in articulation and
naturalness, but frequency limitation is essentially an economic one,
subject to change as conditions change. Recent work on the new
multiple channel carrier systems has indicated justification in these
systems for providing a somewhat wider band, from about 150 to
about 3,500 cycles. ^^
Overall Frequency Response
In describing the frequency characteristic of a transmission system
it has become customary to refer to it as more or less "flat," where
"flat" is assumed to be synonymous with "perfect" as far as the
relative transmission of various frequencies is concerned. In meas-
urements of the elements of an electrical circuit, from which this
terminology came, the word is useful since, when the measurements
are properly made, at any rate, the basis of comparison implied by
the word "flat" is generally understood. This is also true, although
probably to a more limited extent than is generally realized, when the
term is applied to electro-acoustic transmission systems, where free
progressive, plane air waves of various frequencies are transferred to
an electrical system, or vice versa, by means of microphones or loud
speakers.
In the case of a telephone system, however, where a transmitter is
placed close to the lips, and a receiver directly to one ear, and where
the air waves are not free progressive, or plane, use of the word "flat"
implies a basis of comparison which is not self-evident. Much effort
has been given recently to establishing an appropriate reference system,
sufficiently simple in concept and ease of specification, to be useful in
this connection. The result of this work has been a reference telephone
374
BELL SYSTEM TECHNICAL JOURNAL
system which, when spoken into, would give the listener in all respects
essentially the identical sensation he receives in one ear when facing
the speaker directly, with an air path one meter long between the
speaker's mouth and the listener's ear, in surroundings without re-
verberation or noise. Such a reference transmission system has tenta-
tively been called an "orthotelephonic" system.
20
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9-DEClBEL 900-OHM TRUNK
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FREQUENCY IN KILOCYCLES PER SECOND
3.0
3.5
Fig. 7 — Overall orthotelephonic frequency response of
Typical Telephone Connections.
The point of interest here is that when measured by any suitable
objective method, the frequency characteristic of this "orthotele-
phonic" telephone reference system is not "fiat" by a considerable
amount in the ordinarily accepted usage of the term. This departure
from "flatness" is caused by such factors as the frequency directive
1
TRANSMISSION FEATURES OF NEW TELEPHONE SETS 375
characteristic of the mouth, cavity resonance of the ear, and disturb-
ance of the sound field by the head. The individual contribution of
some of these factors is not as yet definitely determined.
Furthermore, for reasons mentioned, it is not self-evident that a
practical telephone system of limited frequency range should be "flat"
with respect to the corresponding frequency band in this more or less
basic orthotelephonic system which is not limited in frequency range.
Having decided on the band width that is desirable and justifiable, it
must still be determined, therefore, what particular frequency char-
acteristics are preferable in this band.
In selecting from the many possible choices, the particular frequency
response that seems best, several factors must be taken into account.
This has been done by a study (under the conditions of actual service)
of the relative results of several different experimental instrument
designs, varying in frequency characteristics. The overall frequency
characteristics of the resultant choice are indicated for two typical
circuit conditions in Fig. 7. These measurements were made with
the artificial mouth and ear ^^ and are plotted with reference to corre-
sponding measurements on an orthotelephonic reference telephone
system. For comparison, the results of similar tests of the earlier
Bell System handset ^^ are shown also.
In considering these overall telephone system frequency response
characteristics in the light of previous discussion, there are several
points of interest:
1. The large increase in response at both higher and lower frequencies
with respect to the older handset, which in itself was a notable
advance in this respect over previous types. This increase
amounts to 10 db or more from about 200 to 500 cycles and
from about 1,700 to 3,000 cycles. This wider frequency range
gives better naturalness of reproduction.
2. The type of the response. The general uniformity and absence of
any marked resonance or irregularity is obvious. For either
average or long loops the entire band from about 300 to
over 3,000 cycles lies within a range of 15 db. It will be noted,
however, that, for the average condition, the response at the
higher frequencies (1,500-3,000 cycles) is distinctly above that
for the frequencies below 1,500 cycles. This characteristic aids
materially in the understanding of the low intensity consonant
sounds. The response on the longer loops would undoubtedly
be correspondingly better if the high frequencies were raised
so that the overall characteristic more nearly resembled that
for the average condition shown. It should be remembered,
376
BELL SYSTEM TECHNICAL JOURNAL
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zz' of high speed pictures of the type represented
by C resulted in strengthening of the reenforcement of the handle at certain
points which reduced the possibility of breakage under normal use conditions.
4000 pictures per second.
E — High speed photography is applied extensively to the study of explosion of gases
in motors, to ballistic problems associated with the explosion of gun powder
and to other rapid phenomena of a self luminous type. This series of pictures
shows the melting and burning of fuse wire under heav\- current conditions.
20-ampere fuse wire is shown during burn-out on direct short. The violence and
extent of the action are well shown in these pictures. 4000 pictures per second.
F — Certain normally isotropic transparent materials become birefringent when
examined in a stressed condition under polarized light. Extended use of this
effect is made in the study of stress distributions in engineering structures and
in models of mechanical parts. High speed photography is now applied to
these photoelastic effects exhibited in a ghptol sample under impact stress
condition. This series of pictures shows impact testing of an unnotched glyptol
specimen in plane polarized light. 300 pictures per second.
G — Poor contact conditions in relays may give rise to improper circuit operation.
High speed motion pictures have been useful in the stud\' of contact chatter in
rela\s and other similar devices. This series of pictures shows normal operation
of contacts. 2000 pictures per second.
H — This series of pictures exhibits contact chatter. In the first picture of this
series the movable contact spring is shown contacting the left fi.xed contacts.
In the second and third pictures, the movable contact spring has been drawn
against the right hand fixed contacts. At this point the current is cut off and
the movable contact springs return to normal as shown in the sixth picture.
Chatter occurs at this point with the movable spring returning to make contact
with the stationary contacts shown at the right. Two cycles of chatter condition
are shown. 2000 pictures per second.
I — This series of pictures shows the No. 14 teletypewriter locking arm lever and
cam during overthrow which gives rise to noisy operation. They illustrate a
topical source of objectionable noise in apparatus of this type. Excessive clear-
ance between the cam and the locking arm lever is shown which results in impact
noise on the return of the locking arm lever. 1800 pictures per second.
J — This series of pictures shows a modified No. 14 telet\pewriter locking arm lever
and cam in which the overthrow has been eliminated with subsecjuent reduction
in noise. It can be seen that the lever arm now closely follows the contour of
the cam. 1800 pictures per second.
K — A knowledge of the fundamentals of speech and hearing is important to designers
of telephone apparatus. High speed motion picture photography has been
applied to problems associated with the production of speech by the vocal
mechanism. The pictures show vocal cords vibrating in production of speech
sound at a frequency of 120 cycles per second. Pictures of this tjpe offer a
unique and practical means of securing nmch useful information relating to the
production of speech. 4000 pictures per second.
L — At L is shown the action of the clapper striking one gong of an experimental
20-cycle ringer. This picture revealed more strokes of the clapper per second
of operation than was desired. This condition resulted in a peculiar acoustic
effect, readily explained from this series of pictures. 2000 pictures per second.
402
BELL SYSTEM TECHNICAL JOURNAL
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HIGH SPEED MOTION PICTURE PHOTOGRAPHY
403
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404 BELL SYSTEM TECHNICAL JOURNAL
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HIGH SPEED MOTION PICTURE PHOTOGRAPHY 405
Bibliography
1. "Motion Picture Camera Taking 3200 Pictures per Second," C. F. Jenkins,
Trans. Soc. Motion Picture Engineers, No. 17, Oct. 1923, p. 77.
2. "The Chronoteine Camera," C. F. Jenkins, /owr. 5. A. E., No. 22, Feb. 1928,
p. 200-2; Trans. Soc. Motion Picture Engineers, No. 25, Sept. 1926, p. 25.
3. "New High Speed Kinematographic Camera," T. Suhara, Proc. Imp. Acad.,
Tokio, No. 5, Oct. 1929, p. 334-7.
4. "New Ultra-Speed Kinematographic Camera," T. Suhara, Report No. 60, Tokio
Univ. Aeronautical Res. Inst., May 1930, p. 187-94.
5. "A Non-intermittent High-Speed 16 M.M. Camera," F. E. Tuttle, Jour. Soc.
Motion Picture Engineers, No. 21, Dec. 1933, p. 474.
6. "The N. A. C. A. Photographic Apparatus for Studying Fuel Spray From Oil
Engine Injection Valves and Test Results From Several Researches," E. G.
Beardsley, N. A. C. A. Tech. Report No. 247, 1927.
7. "The N. A. C. A. Apparatus For Studying The Formation and Combustion of
Fuel Sprays and the Results From Preliminary Tests," A. M. Rothrock,
N. A. C. A. Tech. Report No. 429, 1932.
8. "Stroboscopic and Slow Motion Pictures by Means of Intermittent Light," H.
E. Edgerton, Jour. Soc. Motion Picture Engineers, No. 3, 1932, p. 356.
9. "The Mercury Arc as an Actinic Stroboscopic Light Source," H. E. Edgerton
and K. J. Germeshausen, Rev. of Scientific Instruments, Oct. 1932, p. 535.
10. "High Speed Motion Pictures," H. E. Edgerton, Electrical Engineering, Feb.
1935, Vol. 54, p. 149-153.
11. "Precision Timing of Athletic and other Sporting Events," C. H. Fetter and H.
M. Stoller, Electrical Engineering, June 1933, Vol. 52. p. 386-391.
12. "Time Microscope for High Speed Machines," J. P. Maxfield, Modern Packaging,
July 1936, Vol. 9, No. 11.
13. "High Speed Motion Pictures of the Vocal Cords," W. Herriott and D. W.
Farnsworth, Radiography and Clinical Photography, June 1938, Vol 14, No. 3,
pp. 26-27.
An Optical Harmonic Analyzer *
By H. C. MONTGOMERY
An instrument which makes a Fourier Series Analysis of a func-
tion by optical means has recently been completed. The function
to be analyzed is supplied in the form of a variation in the density or
in the width of the transparent portion of a photographic film. The
analysis is performed by a direct evaluation of the integrals which
form the coefhcients in a Fourier Series, and the results are theo-
retically exact in the sense that the measurement of each harmonic
is independent of the other harmonics which may be present in the
function. The operation of the instrument is largely automatic,
and is rapid enough so that 30 harmonics can be measured in
about a minute and a half.^
APERIODIC function can be represented for all values of the var-
iable by a Fourier Series. A function which is not periodic can
be so represented between any finite limits, although the series may
be entirely unlike the function beyond these limits. If a function is
approximately periodic, the Fourier Series representing adjacent
portions of it will generally be approximately alike.
Although in general an infinite number of terms is required to
represent a function exactly, it is common experience that a great
many functions of practical interest can be closely approximated by a
series of from ten to thirty terms. -
Principle of Operation
The principle of this analyzer was suggested by E. C. Wente of
these Laboratories.^ It may be outlined as follows.
The Fourier Series expansion of a function is given by either of the
following equivalent expressions.^
* Presented at Meeting of Acoustical Society of America, Washington, D. C,
May 3, 1938.
' For comparison, analysis to 30 harmonics on the Henrici type instrument re-
quires five or six hours. A resonance analyzer, such as the vibrating reed type, can
complete an analysis in a few seconds, but the phases will not be given, and if the
function is provided in graphical form it must be converted into an electrical or
acoustic wave form repeated enough times for the resonant elements to reach a steady
state response.
- A description of a number of the more important methods of harmonic analysis,
together with a bibliography, is contained in "Sound Analysis," H. H. Hall, Jour.
Aeons. Sac. Amer., vol. 8, pp. 257-262, April 1937.
3 U. S. Patent No. 2,098,326.
* The expressions in this form apply when the fundamental period is lir. There is
no loss of generality, as the scale of abscissa can always be so chosen as to conform
406
AN OPTICAL HARMONIC ANALYZER 407
fix) = ao + X! «n cos nx -\- J^ bn sin nx (ij
= Co + H Cn cos (nx — (f)n). (2)
1
Comparison of (1) and (2) gives the following relations between the
coefficients in the two forms of the expression :
dn = Cn COS 0„, bn = C„ sin (/)„. (3)
Cn' = a,r + bn', (fin = tan~' -- . (4)
a„
The form (2) giving amplitude and phase angle of the harmonics is
generally more useful, but most methods of analysis give the coefficients
in form (1) necessitating the computation of the amplitude and phase
angle from the relations (4). One of the advantages of the optical
analyzer is that it will give either set of coefficients directly.
The coefficients in the Fourier Series can be determined from the
following expressions: ^
1 r'-'^ 1 r'""
On = - I fix) COS nxdx, bn—~ \ fix) sin nxdx. (5)
Cn = - I fix) COS inx — (j)n)dx, I fix) sin iiix — 4>n)dx = 0. (6)
^Jo Jo
1 r~'^
oo = Co = 2~ I fi^')dx. (7)
We will now describe two methods by which a function can be
represented on a photographic film. In a variable area record the
film, which is elsewhere opaque, contains a transparent portion whose
width at any point is proportional to the function. Such a record is
shown in the upper part of Fig. 1. In a variable density record the
to this requirement. In fact this selection of a proper scale corresponds directly to a
necessary- adjustment of the analyzer.
^ The expressions given in (6) can be derived from the more familiar expressions
(5) as follows. From (3)
a„ cos <^„ + b„ sin (p„ = c„(cos- t^,, + sin^ <>„) = c„,
b„ cos 4>n — a„ sin <^„ = 0.
Substituting values of a„ and b,, given by (5) in these expressions leads at once to the
expressions (6).
408
BELL SYSTEM TECHNICAL JOURNAL
function is represented by gradations in the density of the film such
that the Hght transmission at any point is proportional to the function,
the density being uniform in a direction perpendicular to the axis.
Such a record is shown schematically in the lower part of Fig. 1. With
either type of representation of a function g{x), it will be seen that the
amount of light transmitted through a narrow vertical strip of width
dx is proportional to g{x)dx. If two or more such records are super-
imposed, the light transmitted through all cf them will be proportional
to the product of the recorded functions, provided not more than one
of the records is of the variable area type.
Fig. 1 — Representation of /(.t) and cos nx on film.
The determination of a„ and &„ is now very straight-forward.
Suppose we have fix) recorded on one film and cos nx on another.
For illustration we will assume that f{x) is recorded in variable area
and cos nx in variable density, as shown in Fig. 1, although the only
necessary requirement is that they shall not both be variable area.
If the two films are superimposed, the amount of light transmitted
through both of them between the limits zero and 27r is just the first
integral in (5) and hence proportional to an. Similarly, if the cosine
screen is moved a quarter wave-length along the axis it becomes sin nx
and we have at once bn- If the cosine screen is moved a whole wave-
length along the axis, the transmitted light will go through a maximum.
It will be shown below that this maximum value is proportional to c„,
and the position of the cosine screen at which it occurs is 0„.
One more matter needs to be considered before we write down the
expressions which describe the operation of the analyzer. Since f{x)
^iV OPTICAL HARMONIC ANALYZER 409
and cos nx will in general have both negative and positive values they
cannot be directly represented by the transmission of light, which is
essentially positive. However, the addition of a constant to each
function will eliminate this difficulty, and merely results in a constant
in the measured amplitude, as shown below.
The optical transmission of the film on which f{x) is recorded may
be written
A +/(x),
where A \s a constant large enough to make the expression positive
for all values of x. Similarly, the transmission of the cosine screens
may be written
5n[l + Mn cos {nx - 0)],
where ^ is a parameter denoting the position of the cosine screen along
the X-axis, Mn is a constant somewhat less than unity, known as the
modulation of the record, and J5„ is a constant which is seen to be
the average optical transmission of the screen.
If one or both of these records is cf the variable density type, the
total transmission when they are superimposed will be
T = { B,,IA +/(x)][l + Mn cos {nx - d)yx
= I ABndx + I BrJ{x)dx + | ABnMn COS {nx - d)dx
Jo Jo Jo
BnMJ{x) COS l{nx - (pn) - {0 - 0„)](/x,
0
= 2wBn{A + Co) + TrB„AfnCn COS {d - 0„). (8)
To obtain a„, we take the difference in T for 6 = 0 and 0 = w, which is
seen to be
lirBnMnCn COS 0„ = 27r5,J/„a„. (9a)
iDtam On, we make ^ = -^ and t) =
difference in T
Similarly, to obtain hn, we make ^ = -^ and 6 = -■- , giving for the
lirBnMnCn slu 0„ = lirBnMnbn- (9b)
To obtain c„ and 0„ note that the maximum value of T occurs at
6 = (j)n and the minimum at 0 = 0„ + tt, which serves to determine >„.
The difference between the maximum and minimum values of T is
lirBnMnCn. (10)
410 BELL SYSTEM TECHNICAL JOURNAL
If the factors Bn and ikf„ can be made approximately constant for
all the screens, the coefficients for either form of the Fourier Series are
directly proportional to the change in the amount of transmitted light
for specified pairs of positions of the cosine screens.
Description of the Instrument
The process which the analyzer is required to carry out consists of
superimposing the function to be analyzed on a cosine screen and
measuring the variation in the transmitted light when the cosine
screen is moved along the x-axis. This is repeated with a different
cosine screen for each harmonic which it is desired to measure.
A schematic diagram of the instrument is shown in Fig. 2. The film
containing /(x) is placed in a holder at A and strongly illuminated by
Fig. 2 — Diagram of optical sy.stem.
an incandescent lamp and condensing lens. An enlarged image of
f{x) is formed at i? on a window bounded by two knife edges .750
inch apart. Functions of different length are accommodated by
adjusting the optical enlargement so that the image of the portion of
f{x) to be analyzed will just fill the window. The cosine screens slide
in a track directly behind the window, and receive the image of /(x).
The transmitted light is collected by another lens and brought to a
photocell.
A series of cams and levers is arranged to bring the cosine screens
out of a drum shaped magazine in which they are stored into the
optical path, give them the small motion required for analysis, and
return them to the magazine, which is then rotated to bring the next
screen into position. These operations are all automatic, and the
attention of the operator is required only for the adjustment of the
enlargement and focus and resetting of the cams at the beginning of
each analysis. A photograph of the instrument is shown in Fig. 3.
The variations in the photocell output take place at the rate of about
two cycles per second. These are recorded on a moving chart by an
instrument similar to a high speed level recorder,^ differing from il
chiefly in having a linear instead of a logarithmic scale.
^ "A High Speed Level Recorder," Wente, Bedell and Swartzel, Jour. Acous. Soc.
Amer., vol. 6, p. 121, January 1935.
AN OPTICAL HARMONIC ANALYZER
411
The present instrument has been designed to take records of f{x)
which are from one-sixteenth to five-sixteenths of an inch long and no
higher than their length. The focal length of the enlarging lens is
1.5 inches. The collecting lens is placed quite close to the cosine
screens, and forms an image of the enlarging lens on the plate of the
photocell. With this arrangement the patterns of both f{x) and the
cosine screens are well diffused on the photocell plate, so that surface
variations in sensitivity of the plate are unimportant. The illumina-
tion is uniform across the field to ±2 per cent.
Fig. 3 — The optical harmonic analyzer.
The cosine screens w^ere made on photographic plates, by printing
from variable density motion picture sound track negatives containing
records of pure single frequencies. The pattern thus produced is
about 1X2 inches. The increase in width of the track from about
one-tenth of an inch in the negative to one inch on the plate was
secured by making a "contact" print with negative and plate slightly
separated, and moving the plate sideways under the negative while
printing.
The important requirements for the screens are good wave form,
412 BELL SYSTEM TECHNICAL JOURNAL
uniformity in modulation and average transmission, and accuracy in
wave-length. In the present instrument it was found possible to keep
the harmonic content of the screens down to 5 per cent. The modula-
tion varied from 79 per cent to 94 per cent in different screens, and the
average transmission from 20 per cent to 24 per cent. Variations in
the wave-length of the screens amounted to about 1 per cent.''
It is convenient though not necessary to have good wave form in
the screens. When readings are made in pairs as described above,
the effects of even order harmonics in the cosine screens cancel out.
Moreover, G. R. Stibitz has shown ^ that the cosine screens may have
practically any wave form (not even necessarily periodic), and cor-
rection factors can be derived for them. The process of correction is
rather cumbersome, however, since the correction for each harmonic is
not a constant, but depends on other harmonics present in the function.
Uses of the Analyzer
This instrument was designed particularly to accommodate sound
records on film as used in commercial motion picture work. How-
ever, functions from any other source can be analyzed equally well
if they are reproduced with the proper dimensions "on film. Provision
is made for measurement of the first 30 harmonics. As stated above,
the function must be between 1/16 and 5/16 inch in length and no
higher than its length. At the speeds customarily used for recording
sound on film this corresponds to a fundamental frequency of from
65 to 310 cycles per second, or 1950 to 9300 cycles per second for the
30th harmonic.
The smallest harmonic which the instrument will indicate is about
2 per cent of the peak which it can accommodate. In connection with
this statement, it should be remembered, however, that for many
functions the largest harmonic in the analysis is considerably less than
the peak amplitude of the function, which reduces the effective ampli-
tude range.
An interesting check on the operation of the analyzer can be ob-
tained by making an analysis of a simple geometric wave form.
For example, a single cycle of a saw-toothed wave can easily be
formed by placing a straight edge obliquely across the sound track
slot of the analyzer. Such a wave form is shown at the top of Fig. 4.
It is known that this wave form can be resolved into a series of har-
^ Since a 1 per cent error in wave-length amounts to an error of about one-third
of a wave in the total length of a screen of the 30th order, errors of this magnitude are
quite objectionable in the higher order screens, although unimportant in the low
orders.
8 Unpublished work.
AN OPTICAL HARMONIC ANALYZER
413
9 3
2 -
20
25
Fig. 4 — Analysis of saw-toothed wave.
monies whose amplitude falls off as l/n where n is the order of the
harmonic. The values obtained with this analyzer are shown in the
upper graph in the figure. In the lower graph each harmonic has been
multiplied by n, which should make all the ordinates equal if the
analysis were exactly correct.
The use of the analyzer for the sounds of speech is illustrated in
Fig. 5, which shows the analyses of portions of two vowel sounds made
414
BELL SYSTEM TECHNICAL JOURNAL
VOWEL *'er"
HENRICl ANALYSIS
I, ,l,l,lllll..fllll.|lllk
T I I \ 1 ill
260 300 400 500 1000 2000 3000 4000
FREQUENCY IN CYCLES PER SECOND
100
VOWEL *'0U"
1
OPTICAL ANALYSIS
0
1 .
. 1 i 1 1
hill
II1....1111.1
70 100
"T" — T — I — I — I — i r
200 300 400 500 1000
FREQUENCY IN CYCLES PER SECOND
Fig. 5 — Analyses of spoken vowel sounds.
2000 3000
AN OPTICAL HARMONIC ANALYZER 415
with the optical analyzer. Each is compared with an analysis of the
identical wave form made on a Henrici type analyzer at the State
University of lowa.^ The first sound is a portion of the er in
father. There is a very prominent fourth harmonic, indicating a
strong resonance in the voice at 530 cycles. Other smaller peaks
occur at 1400, 2650 and 3500 cycles. The second sound is a portion
of the diphthong ou in out. It shows two peaks of about equal
magnitude, with a suggestion of a third smaller one. The general
features of the analyses by the two methods are seen to be in good
agreement. A series of such analyses throughout the course of a
spoken sound furnishes a fairly complete description of the changes in
resonance, amplitude, and fundamental frequency which are taking
place. Because of its high speed of operation and convenient applica-
tion to records of speech on film, the present form of the optical
analyzer is especially adapted to such a study of the characteristics
of connected speech.
8 "The Henrici Harmonic Analyzer," D. C. Miller, Jour. Franklin Inst., vol. 185,
pp. 285-322 (1916).
Magnetic Shielding of Transformers at Audio Frequencies
By W. G. GUSTAFSON
The first part of this article is a descriptive discussion of mag-
netic shielding in general. Formulae are then given for the
calculation of shielding efficiency of cylindrical shells for steady
and alternating magnetic fields. By means of these formulae
the shielding efficiency for various types of cylindrical shields has
been calculated for a steady magnetic field.
The second part of the article contains experimental information
on various types of transformer shields. This information supple-
ments the theory in connection with factors which would be very
laborious to treat theoretically.
The theory and the experimental data are coordinated in such
a manner that the shielding efficiency of a particular shield can
be calculated w'ith an accuracy which is sufficient for practical
purposes.
IN connection with the development of repeaters for long distance
telephone lines it is found that noise is introduced into the telephone
lines due to magnetic pick-up by transformers and coils in the repeaters.
This applies also to sound pictures equipment, public and private
address systems, etc., where high gain amplifiers are used. The
stray magnetic field causing this pick-up may be produced by neigh-
boring generators, transformers, rectifiers and other power equipment.
It may also be produced by other amplifier transformers and coils or
by relays located in the vicinity of the disturbed coil. The intensity
of the disturbing field may frequently be of the order of 0.1 oersted
at the point of pick-up. However, a field intensity of the order of
0.02 oersted often causes objectionable noise and under extreme
conditions values as low as 0.001 oersted may be undesirable. As the
gain of the amplifier increases and the demand for good quality
becomes greater, it becomes increasingly important to control magnetic
pick-up. The limiting of this pick-up is in fact today one of the
important problems to be considered in the design of high-gain
amplifiers.
One method by which the magnetic pick-up can be decreased is by
arranging the core structure and winding distribution of the trans-
former in such a way that the voltages induced by an external field
are at least partially neutralized. In many cases, however, this
416
MAGNETIC SHIELDING OF TRANSFORMERS 417
impairs other important characteristics of the transformer and is
therefore undesirable.
Another method is by shielding the transformer from the disturbing
magnetic field. It is the object of this paper to consider such shielding
and to present some data in this connection that may be of general
interest.
Theory
When a transformer is placed in an a.c. magnetic field, there will,
in general, be a voltage induced in the windings. This voltage is
proportional to the intensity of the magnetic field. Therefore, if the
intensity of the magnetic field in the space occupied by the transformer
is reduced, the induced voltage will be correspondingly reduced.
This can be accomplished by enclosing the transformer in a case made
of material which shields against magnetic flux. Let Hi be the
intensity of the field inside the case and He the intensity of the field
when the case is removed. The ratio He/Hi will then indicate the
shielding efficiency of the case. Expressed in decibels:
Shielding efficiency — 20 \ogio He/Hi. (1)
The shielding efficiency of the case depends primarily upon the
permeability and conductivity of the material, and the mechanical
construction of the case.
A high permeability material provides a magnetic path in the walls
of the case of much less reluctance than the air space inside the case.
The greater part of the flux will, therefore, follow the low reluctance
path, and only a small part will enter the space inside the case. The
higher the permeability is, the less the flux that will enter the space
inside the case.* With a steady magnetic field all the shielding is
due to this cause.
An alternating magnetic flux induces eddy currents in the material
of the case as shown in Fig. 1. These eddy currents are a function
of the conductivity and permeability of the material. They may
increase or decrease the shielding efficiency of the case. That is,
the eddy currents iei (Fig. 1), which are due to the component of the
magnetic field perpendicular to a wall of the case, will set up a counter
mmf. which will oppose flux entering the case. In a copper case,
the shielding is primarily due to such eddy currents. On the other
hand, the eddy currents ie2 (Fig. 1), which are due to the component
of the field parallel to a wall of the case will set up a counter mmf.
* It is assumed here, of course, that the source of the magnetic flux is at some
distance so that the amount of flux leaving the source is not appreciably affected
by the case.
418
BELL SYSTEM TECHNICAL JOURNAL
which will oppose the flux following the low reluctance path in the
walls of the case, or what is the same thing, decrease the effective
permeability of this path and will, in that way, decrease the shielding
efficiency. In a case made of high permeability material the latter
eddy currents, i\?, obviously should be reduced as much as possible.
Fig. 1 — Eddy currents produced in a shield by an alternating magnetic field.
If the resistivity of the material is increased, both sets of eddy currents
will be decreased. If, however, the material is laminated with the
sheets parallel to the wall of the case as indicated by section A'-A"
in Fig. 1, the undesired eddy currents will be reduced without affecting
those which are beneficial.
The relative effectiveness of the low reluctance path and eddy
currents in securing good shielding efficiency against magnetic fields
depends mainly upon the frequency of the magnetic field. As a
general rule, we can say that at low frequencies, the effect of the low
reluctance path predominates, while the shielding effect of the eddy
currents, ie\, increases as the frequency increases.
This way of looking at the effect of permeability and conductivity
in a magnetic shield is intended to be purely descriptive and probably
would not be practicable for a mathematical treatment. It is,
however, very suggestive to the design engineer.
As an illustration of the way in which the mechanical construction
of the case may affect the shielding efficiency it is at once clear that
MAGNETIC SHIELDING OF TRANSFORMERS 419
the ratio between the reluctance of the magnetic path in the walls
around the case and the path through the interior of the case depends
upon the size of the case. Also, any openings in the case will obviously
affect its shielding efficiency.
When a magnetic transformer core is placed inside the case, the
reluctance through the interior of the case will decrease and the
shielding efficiency of the case will also decrease. It is therefore
evident that with one type of magnetic core a case might have a
different shielding efficiency than with another.
To obtain general mathematical relations between the shielding
efficiency and the various factors mentioned above is very difficult.
By making some simplifying assumptions, however, relations can be
obtained that will be useful from a practical standpoint, although
they will necessarily be somewhat limited in application.
A great deal of work on the shielding efficiency of shields constructed
of different magnetic materials has been done by various investigators.*
Except in a few of the more recent papers, f consideration has been
restricted to a steady, uniform, magnetic field where no eddy currents
are produced and where, therefore, the shielding is due entirely to the
magnetic properties of the material. They have also usually limited
themselves to spherical and cylindrical shields. The cylinders have
been considered of infinite length with the direction of the disturbing
magnetic field perpendicular to the axis of the cylinder. However,
with cylinders of finite length they have found that for moderate
shielding efficiencies, at points inside a cylinder at a distance from
the end equal to its diameter the shielding efficiency is approximately
that of an infinite cylinder. The ratio between the shielding efficiency
of a cylinder and that of a sphere, the radii of the two, the permeability,
the thickness and construction of the walls being the same, varies
from approximately 4 : 3 in favor of the sphere for very thin shells
to 9 : 8 in favor of the cylinder for very thick shells. This gives some
idea of how the shape of the shield affects the shielding efficiency.
Investigations by the various investigators referred to above show
that the shielding efficiency of two or more concentric cylinders or
spheres may be vastly greater than that of one cylinder or sphere,
the amount of magnetic material being the same. They have given
mathematical relations between the shielding efficiency, the permea-
bility and the mechanical dimensions of both spheres and cylinders.
Although these relations have been derived for a steady magnetic
field, they may also be applied with certain limitations to an alternating
* See Bibliography.
t The most important exceptions are articles No. 18, 19, 21, and 28 in the
Bibliography.
420 BELL SYSTEM TECHNICAL JOURNAL
magnetic field. The permeability used in this case will, of course,
be the effective permeability for the particular conditions and fre-
quency under consideration. These relations refer only to that
portion of the shielding effect which is independent of the eddy
currents ie\ (Fig. 1), and the total shielding effect will, in general,
be somewhat greater.
In an article in the Physical Review of October, 1899, A. P. Wills
considers the cases of three concentric cylinders and spheres. Due
to the fact that spherical shields are less suited for our purpose I
will give the equations for cylindrical shields only. Wills' formula for
three cylinders for large values of permeability is given by the following
equation :
g = 1/4 /i { (1 — qiq^qz) + 1/16 ix^fiifinninnna
+ 1/4 ju[(WlW3 + W1W2 — «l«2W3)Wl2
+ («1«3 + «2W3 — WiW2«3)W23 — W1W3W12W23] } +1. (2)
In this equation
. = |, (3)
where He is the density of the magnetic field at a point P with the
shield removed and Hi is the density of the magnetic field at that
point when enclosed by the shield, fj, is the permeability of the
material at the frequency in question. We have
qi = nV-Rl^ wi = 1 - gi,
52 = ^2V^2^ «2 = 1 - §2, **
53 = rs''IR,\ n,= I - §3, (4)
212 = -RlV^2^ W12 = 1 — §12,
§23 = Ri^lri, fiiz = \ — q23,
where n, Ri, r^, etc., are the various radii of the cylinders as shown
by Fig. 2.
By making qz = \ (or Uz = 0), in (2), we get the relation for two
concentric cylinders.
g = 1/4 /i(l - qiq2 + l/Annifiifin) + 1. (5)
By making 52 = 1 (or W2 = 0), (5) changes into an equation for one
shell only
g = 1/4 Ml -g) + 1. (6)
MAGNETIC SHIELDING OF TRANSFORMERS
421
It has been shown (A. P. Wills, Phys. Rev., vol. 24, page 243,
February 1907) that for a shield of predetermined size, that is when
the smallest and the largest radii (ri and R^ in the case of three cylin-
ders. Fig. 2) are specified, the radii of the surfaces of the successive
I
Fig. 2.
cylinders should be in a geometric progression to give the most efficient
shield. That is, we should have qi — qi — qz = qn = §23 = q- Equa-
tion (2) then becomes
g = 1/4 m(1 - g' + 1/16 fx^n' -f M«' + 3/4 m«') + 1. (7)
For two cylinders we get
g = 1/4 m(1 -2^ + 1/4 M«^) + 1. (8)
In these equations, the following relations hold between n and Rs
for (7) and ri and R2 for (8)
Rs = rJ^[i^ R, = rif^\ (9)
The effect upon the shielding efficiency of varying 5 (Fig. 3) from
422
BELL SYSTEM TECHNICAL JOURNAL
zero to P keeping rijRi = r^jRi can be obtained by means of the
following equation
g- 1/4m
where
1 -^ + 1/4m(i --^Xn,,
912 \ Vgi2/
" R2
+ 1>
(10)
If Ri/r2 (that is Vgi2) is varied from 1 to R^/ri the desired result is
obtained.
Assume in Fig. 3 the thickness of the two cylinders to be the same,
that is, R\ — r\ = R2 — ^2- The variation in shielding efficiency vs. 5
FiR. 3.
or Vgi2, is then given by equation (5), with §2 expressed in terms of
512 and q\ as follows :
Vg2 =
1 + Vgi2[l - Vgi]
(11)
In an article in the Philosophical Magazine of February 1933 L. V.
King has developed relations for the shielding efficiency of spherical
and cylindrical shells taking into account the effect of induced currents.
The following equations for an infinitely long metallic cylinder have
been picked from his paper. For a non-magnetic shell, the thickness
of which is small compared to its radius, the shielding ratio, g, is
given by
g = 1 cosh {ka) + 1/2 ka sinh {kd) | , (12)
»
MAGNETIC SHIELDING OF TRANSFORMERS 423
where a is the radius and d the thickness in cm. k is given by
where/ is frequency in cycles and p is resistivity in ohms per centimeter
cube. At low frequencies (12) reduces to
1+,-M^io-
(14)
which is good up to about 10'^ cycles. The direction of the disturbing
magnetic field in (12) and (14) has been assumed perpendicular to
the axis of the cylinder.
Other formulae which take into account both conductivity and
permeability are also given in King's article. They are, however,
rather complicated and require elaborate calculations.
Mr. S. A. Schelkunoff in an article in the October 1934 issue of the
Bell System Technical Journal has derived formulae which are com-
paratively simple although they take into account both conductivity
and permeability. His treatment is quite different from that presented
above and his results are expressed in terms of radial impedances.
P'or an infinitely long cylindrical shield the diameter of which is large
compared to the radial thickness of the shield the shielding efficiency,
5, is given by
S^R^A. (15)
In this formula R is the sum of the reflection losses at the surfaces of
the shield. We have
i? = Z 20 logio ■ " t I db, (16)
* where kn is the ratio of the radial impedance in the first medium to
that in the second. That is,
kn=^' (17)
The radial impedance for a good dielectric is given by
Z = lirfixpi ohms. (18)
For a metal
Z=JMl^. (19)
424
BELL SYSTEM TECHNICAL JOURNAL
In (18) and (19) / is the frequency in cycles, p is the radius in cms.,
g is the intrinsic conductance in mhos/cm., and n is the intrinsic
inductance in henries/cm.*
A in (16) is the sum of the attenuation losses in the successive
shells. For any one shell
A = 8.686a/ db,
where a = ^Tgfxf and / is the radial thickness in cms.
(20)
Calculated Curves
By means of the equations (2) to (11) the shielding efficiency of
various types of cylindrical shields has been calculated. The permea-
150
140
130
120
no
)
1
j 100
)
J 90
'' 80
)
r 70
: 60
j
, 50
] 40
\ 30
20
10
0
Fig._4 — Shielding efficiency of one, two, and three concentric cyhnders
(for zero frequency).
bility considered is 5000 which is readily obtainable at low frequencies
and field strengths by means of permalloy .f The calculations are
given in the form of curves in Figs. 4, 5 and 6.
* Thus in empty space (or dielectrics, approximately) n — 4x10"' henries/cm.
In general yu = 47r;uolO~' where mo is the intrinsic permeability referred to empty
space as unity.
t Arnold and Elmen, "Permallo}'," Journal Frankliii Institute, May, 1923,
pp. 621-632.
E3l
cl^
^
^
■
^
/
^
a fCALE 1)
—
—
J
^
€^
M = 5000 R|/r, = Vr I = "Vr2= Vr2= ^Vps
A ONE CYLINDER Rn/r| = R|/p
B TWO CYLINDERS Rn/r|=R2/r,
C THREE CYLINDERS Rn/r|= Rs/r,
CYLINDERS CONSIDERED INFINITELY LONG
/
®
/
c
r
'^si
:2^
T
0 AX
IS Of
" CY
JND
ERS
00
1.04
1.08
1.12
1.16
1.20
1.24
1.28
1.32
1.36 SCALE 1
JO
1.50
1.70
1.90
2.10
2.30
2.50
2.70
2.90
3.10 SCALE 2
MAGNETIC SHIELDING OF TRANSFORMERS
425
The curves of Fig. 4 show the shielding efificiency of one cylinder
and of two and three concentric cylinders with air space between.
The shielding efficiency is given as a function of the ratio between the
outside and inside radii of the shield, that is, i?„/ri, where Rn is the out-
side radius of the outside cylinder and n is the inside radius of the
inside cylinder. These curves show the relative shielding efficiencies
of 1, 2 and 3 cylinders and give numerical values for ^l equal to 5000.
The relative dimensions of the cylinders are such that the ratios
between the inside and the outside radii of the cylinders and of the
air spaces between the cylinders are in geometric progression. These
curves show that when a very high shielding efficiency is desired it is
not only advantageous but necessary to use two or more cylinders.
Thus with one cylinder the maximum shielding efficiency that can be
considered practical when /x = 5000, is approximately 50 db. The
maximum theoretical limit for one, two and three cylinders is 62 db,
124 db and 186 db respectively when n = 5000.
1 1 1 1
T|=Tp = 0.042" INCH
^
^
/
/
T| =
r2=o
.014
INCH
f —
/
/
^
M =
5000^^^^^
/
©'^
0 200 400 600 800 1000 1200 1400 1600 1800 2000
SEPARATION IN MILS (S)
Fig. 5. — Shielding efificiency of two concentric cylinders versus the air-gap between
them. Thickness of the wall of each cylinder kept constant. Zero frequency.
In Fig. 5 is shown the shielding efficiency of two concentric cylinders
vs. the thickness of the air space between the cylinders. The thick-
nesses of the walls of the two cylinders are equal. Two thicknesses
of the walls of the cylinders have been considered, namely, .014" and
.042". An interesting fact is brought out by comparing the curves of
Fig. 5 with curve "5" of Fig. 4. For example, when the air space
is .042", the upper curve of Fig. 5 gives a shielding efficiency of two
cylinders with an air space between them, the thickness of which is
the same as that of the cylinders. The ratio between the outside
\
426
BELL SYSTEM TECHNICAL JOURNAL
radius and the inside radius is 1.097. The shielding efficiency of this
combination is approximately the same as that of two cylinders having
the same ratio Rnlri as given in Fig. 4, curve "5." This shows that
the condition that the radii of the cylinders should be in geometric
progression is not very critical, at least, not for the value of i?„/ri
under consideration.
100
95
90
85
80
uj 65
Q
Z 60
O 55
Z
^
-—
—
P-;
/
y'
°°"ic;^
■^
/
/
^
N
/
^
^^
— "
^2^
\
/
/
^'
^
\
V,
\
/
/
^
^
—
N
v
\
rr-2^
/
/
x"
^-
-^
\
\
/
-^
^
"V
N
\
\
\
//
/
^
N,^
\
\
\
/.
^
^
.^
//
— s
N
\
\
/
^
\^
\
V
\
—
—
^
N
s
\\
\
^53
s^^
^
-^
\
\
\\
"^
\
\
)
\\
N
\
\
\
^^^ !^J ^
i= 5000
\
s\
J
r, = 1 INCH
\
\
\\
^
0 10 20 30 40 50 60 70 80 90 100
SEPARATION IN PER CENT (S/pXIOO)
Fig. 6 — Shielding efficiency of two concentric cylinders versus the air-gap
between them. Overall thickness of the wall of the shield (P) kept constant. Zero
frequency.
The curves of Fig. 6 also show the shielding efficiency of two con-
centric cylinders vs. the air-gap between them. In this case, however,
the total thickness P of the wall of the double cylindrical shield is
kept constant and the air-gap is increased from zero to P at the
expense of the cylinders. When the air-gap is zero, the shielding
efficiency is therefore that of a solid cylinder, the thickness of the
MAGNETIC SHIELDING OF TRANSFORMERS 427
wall of which is P. When the air-gap is equal to P, the thickness of
the two cylinders is zero, and hence the shielding efficiency is zero.
The air-gap is so located that the radii of the two cylinders are in
geometric progression.
Experimental Data
From the discussion under "Theory" it is evident that although the
equations (2 to 11 inch) were derived with the assumption of a steady
magnetic field the above calculated values apply equally well to an
alternating magnetic field if the effective permeability is used. How-
ever, the results are far from sufficient to determine the shielding
efficiency of a magnetic shield for a transformer. The shielding due
to the eddy currents, iei (see Fig. 1), is not included. To include
this the formulae (12) to (20) inclusive must be used. The equations
have been derived with the assumption that the length of the cylinders
is infinite. In practical applications this is obviously not so. Covers,
however, approximately counterbalance the effect of the finite length
of the cylinders. The magnetic core of the transformer also materially
affects the shielding efficiency. In connection with such factors as
these which would be very laborious to treat theoretically some
experimental information will now be given. The frequency range of
the disturbing magnetic field was limited to from 50 to 4000 cycles.
The shielding efficiency has previously been defined as follows:
Shielding Efficiency = 20 logio He/Hi. (1)
From the standpoint of the shielding of transformers we are primarily
interested in the reduction of the transformer terminal voltage which
is caused by the disturbing magnetic field. For this reason it will be
found convenient to define the shielding efficiency in connection with
transformers in decibels as follows :
Shielding Efficiency = 20 logio Ee/Ei, (21)
where Ee is the terminal voltage due to the disturbing magnetic field
with the shield removed and Ei the corresponding voltage with the
transformer inside the shield. In addition, Ee and Ei are restricted
to the maximum terminal voltages, with respect to position, that is,
the transformer is assumed to be in that angular position with respect
to the direction of the magnetic field, in which the maximum terminal
voltage is obtained. With an unshielded shell type transformer, for
example, this position would be that in which the axis of the winding
coincides with the direction of the disturbing magnetic field. This
restriction is necessary for the definition (21) to be of any value.
428
BELL SYSTEM TECHNICAL JOURNAL
With a small air core coil and an infinite cylinder the axis of which is
perpendicular to the axis of the winding and to the direction of the
disturbing magnetic field the two definitions are equivalent.
The circuit used in making measurements consists of a field coil
producing a magnetic field, a pick-up coil and a vacuum tube voltmeter.
The field coil is a loop two feet in diameter consisting of 500 turns of
wire. The magnetic field at the center of this coil is uniform over a
space sufficiently large for the purpose. The pick-up or search coil
when placed in the magnetic field will have a voltage produced across
its terminals. This voltage is measured with the vacuum tube
voltmeter and Eg and Ei in equation (21) are thus obtained. The
size of the shell type core on which the pick-up coil is wound is
3" X 1 29/32" X 1" where the 1 29/32" dimension is parallel to the
axis of the winding, the 1" dimensions being the pile-up of laminations.
Unless otherwise mentioned this coil was used in all of the following
measurements.
Permalloy having an initial permeability of the order of 5000 at
low frequencies was employed for both the core and the shield through-
out this investigation.
Permalloy Cases
In Fig. 7 is shown the shielding efficiency vs. frequency of a rec-
tangular permalloy case which consists of five contiguous layers of
^<(.y^
^
s^
.^^
N
s
V
•\
^
-^
X
-O^^l^
^
,^^
X
\
\
s
">
\
-
^
^
500 1000
FREQUENCY IN CYCLES PER SECOND
Fig. 7 — Observed shielding efficiency of laminated permalloy case.
.014" thick permalloy sheet. The size of this case is approximately
2 3/8" X 2 1/2" X 3 1/4" and the position of the coil inside the case
is such that the axis of the winding is parallel to the 2 1/2" dimension.
The relative size of the coil and case is such that there is approximately
MAGNETIC SHIELDING OF TRANSFORMERS
429
1/8" clearance between the core of the coil and the case. The shielding
efficiency is given for a coil having a permalloy core and also for a
coil having a non-magnetic core. These curves illustrate the effect of
the magnetic core upon the shielding efficiency. At low frequencies
the shielding is mainly due to the magnetic properties of the shield
material. As the frequency increases, however, the effect due to the
eddy currents, iei (Fig. 1), increases and the shielding efficiency
increases. At approximately 300 cycles a maximum is reached and
from here on up to 4000 cycles the shielding efficiency decreases.
This is due to the fact that at these frequencies the eddy currents, ie^
(see Fig. 1), decrease the effective permeability of the material at a
greater rate than the shielding efficiency is increased due to the eddy
currents iei. The slope of the curves at 50 cycles is not zero. This
shows that even at 50 cycles there is a considerable shielding effect
due to the beneficial eddy currents.
40
^
>
35
PERMALLOY
WITH rnvFR
CASE
g
-
-
V
V^,^^
<0
'
_l
(0
■
UJ
Q
PERN/
Al ir
Y
.^
V
NO COVER
z
[ 1
u.
ii.
o IS
Z
Q
4 10
I
5
0
^
^
^^
" ■
-
SILICON STEEL
WITH COVER
.^
^
-
■
^
"
200 500 1000
FREQUENCY IN CYCLES PER SECOND
Fie. 8-
-Observed shielding efficiency of cases made of 1/32" thick permalloy
and siUcon steel sheet.
The curves of Fig. 8 show the shielding efficiency vs. frequency of
a cylindrical permalloy case, the thickness of the walls of which is
1/32". The approximate dimensions of this case are 3 1/4" high
X 2 5/8" diameter and the relative dimensions of the case and core
are such that there is approximately 1/8" clearance between the core
and the case. A comparison between the two curves for permalloy
shows the effect of the cover upon the shielding efficiency.
430
BELL SYSTEM TECHNICAL JOURNAL
A comparison between the curve for permalloy and that for silicon
steel on Fig. 8 gives a striking example of the advantage of using
permalloy instead of steel.
Effect of Air -Gap Between Core and Shield
It has been pointed out previously that the magnetic core decreases
the reluctance through the interior of the case and in that way de-
creases the shielding efficiency. When the size of the core is such as
to almost fill the case, that is, when the air-gap between the core and
the case is small, this effect becomes large. In Fig. 9 are given some
d IN INCHES
363 0.094 0.125 0.156
}t^35
UJ25
O
O
il5
SlO
(□;
A
^IR rr^^
^—~~1....^0R£
^i^MALLOY COR
E _ ^
^
^\^
"^-^ _
-2!!I£RENCE
NO. 3 NO. 4
CYLINDER
Fig. 9 — Observed effect of air-gap between core and shield. Cylindrical shield.
Frequency 70 cycles.
data in this connection for cylindrical shields. Six permalloy cyHnders
were used in this illustration. They are so constructed as to fit over
each other, the smallest fitting snugly over the transformer core
("d" = 0). Each cyUnder consists of two layers of .014" thick
permalloy sheet. The cyHnders have been numbered 1 to 6 from
the smallest to the largest, respectively. Curve "A " of Fig. 9 shows
the shielding efficiency with a non-magnetic core and curve "B"
gives the corresponding information with the permalloy core having a
permeability of approximately 5000 at 70 cycles and low field strengths
which are the conditions under which the measurements were made.
MAGNETIC SHIELDING OF TRANSFORMERS
431
These curves have been drawn discontinuously because the shielding
efficiency is a function not only of "J" but of other factors such as
permeability, size of the cylinders, etc. Curve "C" on the other
hand is primarily a function of "d'' and has, therefore, been drawn
continuously. This curve gives the difference between the shielding
efficiency with an air core and with a permalloy core and shows the
advantage of increasing the air-gap between the core and the cylinder.
Thus, for example, in this particular case approximately 8 db better
shielding efficiency is obtained with an air-gap of 1/16" than if there
is no air-gap.
0.028
d IN INCHES
556 0.084
BOX
Fig. 10 — Observed effect of air-gap between core and shield. Rectangular shield.
Frequency 70 cycles.
Similar curves are given in Fig. 10 for rectangular boxes of the same
height and same wall thickness as the cylinders. The measurements
were made at 70 cycles per second. Due to the larger contact area
between the core and the shield the effect of the air-gap is much
greater here than with cylinders.
As the efifective permeability of the walls of the shield increases
the effect of an air-gap increases, other things being equal. In general
it may also be said that as the shielding efficiency increases (due to
increased thickness of the walls, for example) the effect of an air-gap
increases.
If there is an appreciable air-gap between the core and the shield
a large variation in the effective permeability of the core will affect
432
BELL SYSTEM TECHNICAL JOURNAL
the shielding efficiency very Httle. That is, practically the same
results will be obtained with a silicon steel core, having a permeability
of 400 as with a permalloy core having a permeability of 5000. On
the other hand if the silicon steel core is replaced by an air core the
change in the shielding efficiency will be of such an order as is indicated
by Fig. 7. The reason for the small effect of changing from a per-
malloy core to a silicon steel core as compared to the changing from
a silicon steel core to an air core is, of course, due to the fact that in
the first case there is a decrease in permeability of 12.5 : 1 while in
the second case the corresponding reduction in permeability is 400 : 1.
High Efficiency Shields
A shielding efficiency of from 20 to 50 db is, for many purposes,
sufficient in connection with the shielding of transformers. Occasions
arise, however, when a shielding efficiency much greater is desired.
500 1000
FREQUENCY IN CYCLES PER SECOND
Fig. 11 — Observed shielding efficiency of various high-efficiency shields.
To accomplish this by means of a simple case, magnetic material
having a permeability much greater than is now readily available
would be needed. The curve B of Fig. 11 gives the shielding efficiency
MAGNETIC SHIELDING OF TRANSFORMERS 433
of a permalloy cylinder which has a permeability of approximately
5000. This cylinder is 4.5" high, inside diameter 2.5", and thickness
of wall equal to .07". By increasing the thickness of the wall the
efficiency would be only slightly increased. This is evident from a
study of Fig. 4. However, by placing a second cylinder over "JB" a
substantial improvement is obtained (Curve C). Still greater shielding
efficiency is obtained by placing a third cylinder over the former two
as shown by curve D. The dimensions of the second and third cylin-
ders are such that the ratios between the outside and inside radii of
the three cylinders and of the air-gaps between them are approximately
in geometric progression. The height of the second and third cylinders
is also 4.5" and the effective permeability at low frequencies and field
strengths approximately 5000.
Since the effective permeability of the permalloy used in the above
shields is close to 5000 we can compare these data with the theoretical
curves of Fig. 4, which were calculated with the permeability assumed
equal to 5000. This comparison shows that the theoretical analysis
of the shielding of infinite cylinders against steady magnetic fields
may be applied to the shielding of transformers. Due to such factors
as a magnetic core inside the shield, eddy current shielding, end
effects etc. only an approximate check can be expected. At 50 cycles
per second the measured values for one cylinder and for two and three
concentric cylinders are 40, 64, and 80 db respectively. Corresponding
calculated values as given by Fig. 4 are 41.5, 66, and 89 db respectively.
It is evident from the effect of the copper cylinder between two
permalloy cylinders as shown by curve E (Fig. 11) that three per-
malloy cylinders with copper cylinders between the inner and middle
and between the middle and outer will give a shielding efficiency of
the order of 100 db (voltage ratio EejEi = 10^) from 50 to 4000 cycles.
Effect of Covers
The information given in Fig. 12 shows the importance of covers.
This figure gives the shielding efficiency of a cylinder which consists
of two layers of .014" permalloy sheet. Curve A gives the shielding
efficiency without any covers and curve B shows the advantage of
adding covers which overlap 1/2" and consist of two layers of .014"
permalloy sheet. The two curves C give the shielding efficiency of
the same cylinder provided with flat plate covers, Ci representing
covers .014" thick and C2 covers .028" thick. The relative size of
the coil and cylinder is such that there is a clearance of approximately
1/16" between the core and the magnetic shield. It is evident that
in this particular instance the covers are very important. Regarding
434
BELL SYSTEM TECHNICAL JOURNAL
40
m
U 35
o
z
V 30
o
z
Uj
o
c:25
If) 15
-
A
B
C
— ,
-1
-
====^
■
""^
.^
^-^^
V
^
^
X
^^
B
C2
■
—
-
-
_.^
C|
■^
A
=0 100 500 1000 5000
FREQUENCY IN CYCLES PER SECOND
Fig. 12 — Observed effect of covers on cylindrical shields.
contact between the cover and the cylinder it may be shown that at
low frequencies this is immaterial while at higher frequencies the
reverse is true. However, even at these higher frequencies a small
overlap (as for B) is sufficient.
Examples of the effect of covers upon the shielding efficiency are
also given by Figs. 8 and 11. The efifect of covers on a copper cylinder
as shown by curves A and B (Fig. 13) is of special interest.
40
20
WITH COVERS
^
B
COPPER
CYLINDER
bsS
^--^
A
10
0
NO COVERS
_
50 100 200 500 1000 2000 5000
FREQUENCY IN .CYCLES PER SECOND
Fig. 13 — Observed shielding efficiency of a copper cylinder.
Use of Copper
The curves A and B of Fig. 13 give the shielding efficiency vs.
frequency of a cylinder made of copper. This cylinder has an inside
diameter of 2 5/8" and is 4.5" high. The thickness of the wall is
1/16". The shielding efficiency is here entirely due to the eddy
MAGNETIC SHIELDING OF TRANSFORMERS
435
currents, iei (Fig. 1). Curve B shows that, after the effect due to the
open ends of the cylinder is eliminated by means of covers, the shielding
efficiency is approximately proportional to the logarithm of the
frequency.
Although copper has a very low shielding efficiency at low frequencies
when used alone, tests show that under certain conditions it is very
effective when used in conjunction with permalloy. This is illustrated
by a comparison between the curves C and E of Fig. 11. The copper
cylinder is similar to the one referred to above except that the thickness
of the wall is only 1/32". The permalloy cylinders are those for which
the shielding efficiency is given by curve C of the same figure.
Another striking example of the use of copper in conjunction with
permalloy is furnished by Fig. 14. The curve A gives the observed
S40
COPPER PLATES EACH
30X 4.5'
HIG(-
A d=y32
NO COPPER
d-
rr-ij
CUNijIbIS Ul- d. LATtKi
OF 0.014" PERMALLOY
-d SHEET
SPOOL PERMALLOY CORE
B d = >32 .COPPER PLATES
„ j/32" THICK
C d = !/|6' NO COPPER
„ PLATES
D d = kl6, COPPER PLATES
, „!/|6"THICK
E d = yi6,COPPER PLATES
K?" THICK
IcoreI
PERMALLOY
BOX OPEN AT-^
i_j4
BOTH ENDS
,
>
^
^
^
^
-^
»B,D,
iE
1
y
/
/
X
^
C
-^
■^
A
r —
— -
'
'^
C
A
500 1000
FREQUENCY IN CYCLES PER SECOND
Fig. 14 — Observed effect of copper between core and permalloy shield.
shielding efficiency of a permalloy box when there is an airspace
between the core and the box of 1/32" {d = 1/32"). If 1/32" thick
copper plates are inserted between the core and the box (see Fig. 14),
there is a great improvement in the shielding efficiency as a comparison
between the curves A and B shows. This improvement is 10 db at
50 cycles although the effect of the copper plates alone would be of
the order of one db, as is evident from the curves on Fig. 13. At
higher frequencies the improvement is still better. It is approximately
15 db between 100 and 4000 cycles. Approximately the same results
436 BELL SYSTEM TECHNICAL JOURNAL
are obtained with a spacing of 1/16". A comparison between the
curves C and E shows the improvement which is obtained in this
case with 1/32" copper plates. The effect is somewhat less than with
a spacing of 1/32", at least at frequencies below 1000 cycles. At
low frequencies copper plates, 1/16" thick, show a slight improvement
over the 1/32" copper plates as is shown by the curve D.
Curve B on Fig. 10 shows that if the airspace between the core and
the box is small the shielding efficiency is very low. In a case Hke
this a copper spacer is very effective. For example, a 5 or 10-mil
copper plate replacing an airspace of the same thickness will greatly
improve the shielding efficiency.
General
The foregoing is a discussion of the magnetic shielding of trans-
formers from external magnetic fields. The reverse problem of
shielding a transformer or coil so as to prevent its magnetic field from
affecting other apparatus has not been considered. However, it is
safe to assume that approximately the same degree of shielding will
be obtained, provided the leakage field does not produce excessive
saturation in the shield. That is, assuming that a power transformer
is producing a disturbing magnetic field in the space occupied by an
input transformer, then a shield over the power transformer will
produce approximately the same effect as a shield over the input
transformer, where each shield has been constructed in accordance
with the information on the foregoing pages. This has been demon-
strated experimentally in an article by J. E. R. Constable in the
Wireless World of February 26, 1937.
Although this paper has been restricted to the magnetic shielding
of transformers it is equally applicable to any apparatus which is
susceptible to inductive pick-up. This is because in any apparatus
where there is inductive pick-up there is in effect a coil. It may be
an actual coil and it may be only a loop of lead wires.
The author wishes to thank Mr. E. T. Hoch for many helpful
suggestions.
Bibliography
1. J. Stefan, Wied. Annalen, Vol. 17, p. 928, 1882.
2. A. VV. Rucker, Phil. Mag., Vol. 37, p. 95, 1894.
3. H. DuBois, Wied. Ann., Vol. 63, p. 348, 1897; Vol. 65, p. 1, 1898, also Electrician,
Vol. 40, 1897, pp. 218, 316, 511, etc. cont.
4. A. P. Wills, Phys. Rev., Vol. 9, p. 193, 1899; Vol. 24, p. 243, 1907.
5. James Russell, Roy. Soc, Edhib., Trans., Vol. 40, p. 631, 1903.
6. E. F. Nichols and S. R. Williams, Phys. Rev., Vol. 27, p. 250, 1908.
7. W. Esmarch, Ann. der Phys., Vol. 39, p. 1540, 1912.
8. W. W. Coblens, Bui. Bu. of Standards, Vol. 13, p. 423, 1916.
MAGNETIC SHIELDING OF TRANSFORMERS 437
9. C. Benedicks, Ann. der Phys., Vol. 72, p. 236, 1923.
10. R. H. Barfield, Inst. Elec. Eng., Vol. 62, p. 249, 1924.
11. D. W. Dye, Jour. Set. Inst., Vol. 3, p. 65, 1925.
12. A. V. Hill, Jour. Sci. Inst., Vol. 3, p. 335, 1925.
13. J. H. Morecroft and A. Turner, /. R. E., Proc, Vol. 13, p. 477, 1925.
14. H. Pleijel, Svenska Ingeniorsvetenskapsakademiens Handlingar, NR 49, 1926.
15. H. L. Curtis, A. I. E. E. Journal, Vol. 48, p. 453, 1929.
16. S. L. Gokhale, A. I. E. E. Trans., Vol. 48, p. 1307, 1929.
17. N. Hillers, Telefunken Zeitung, Vol. 13, pp. 13-28, 1932.
18. L. V. King, Phil. Mag., Vol. 15, p. 201, February 1933.
19. W. Lyons, I. R. E., Proc, Vol. 21, p. 574, April 1933.
20. G. W. O. Howe, Wireless Engr. and Exp. Wireless, Vol. 11, p. 347, July 1934.
21. S. A. Schelkunoff, Bell Sys. Tech. Jour., Vol. 13, p. 532, October 1934.
22. Radio Engg., Vol. 15, p. 11, July 1935.
23. T. E. Sterne, Rev. Sci. Inst., Vol. 6, p. 324, October 1935.
24. W. F. Randall, The Nickel Bulletin, Vol. 9, No. 4, p. 73, April 1936.
25. R. Bachstroem, Arch. f. Elektrotechnik, Vol. 30, p. 267, April 1936.
26. S. A. Schelkunoff, Radio World, Vol. 29, p. 47, April 1936.
27. Samuel Levy, I. R. E., Proc, Vol. 24, p. 923, June 1936.
28. C. W. Oatley, Phil. Mag., Vol. 22, p. 445, September 1936.
29. J. E. R. Constable, Wireless World, Vol. 40, p. 198, February 1937.
30. W. Herzog, E. N. T., Vol. 14, p. 81, March 1937.
Coaxial Cable System for Television Transmission*
By M. E. STRIEBY
THE reports which have been made on the progress in television
development increase the expectation that the broadcasting of
visual programs will soon be realized. In anticipation of that result,
the Bell Laboratories has been engaged for some time in the develop-
ment of wire line circuits for transmitting television signals between
studios and broadcasting transmitters, or between cities, as may some
day be required if television follows in the footsteps of sound program
broadcasting.
The wide frequency bands required for television and the dearth
of available frequencies appear to force the broadcasting of television
signals into the ultra-high frequency range. At these high frequencies,
the coverage which can be obtained from a broadcasting station is
very limited as compared to that obtainable in the sound broadcasting
frequency range. Hence, if television programs are to reach large
sections of the country simultaneously, the provision of interconnec-
tions between large numbers of television broadcasting transmitters will
become even more important than it is today for sound broadcasting.
Coaxial cables have received much publicity as transmission lines
for television. The original conception and use of the coaxial form
of cable was first as a low frequency submarine conductor and later
as a lead-in for radio antennas. The idea of a coaxial cable or other
medium for the transmission of very broad frequency bands orig-
inated in the course of telephone development in America.^ The first
lengths of such cable for broad-band transmission were made here and
its first use for the transmission of a large number of simultaneous
telephone conversations was between New York and Philadelphia.^
In this country the important reason for developing coaxial cable
systems was, and still is, that they appear to offer material economies
in the provision of large groups of long distance telephone facilities.
Television has been secondary.
Recently experiments have been made on the transmission of
television signals over the coaxial cable between New York and
Philadelphia. This cable contains two coaxial conductor units within
* Presented at A. I. E. E. Winter Convention, New York City, Jan. 27, 1938.
Published in June 1938 issue of Elec. Engg.
438
COAXIAL CABLE SYSTEM FOR TELEVISION TRANSMISSION 439
a lead sheath about J^^" in diameter as indicated in Fig. 1. Two
coaxial units were provided because, for long distance telephone
operation four-wire operation is preferable, one coaxial being employed
for transmission east to west and the other west to east. Each coaxial
unit is made up of a 13-gauge inner conductor on which hard rubber
disks have been placed at intervals of ^ of an inch. The outer
metallic tube is made up of 9 overlapping copper tapes so designed
that they form essentially a solid copper tube about 20 mils in thickness
and .267" in inside diameter.
The transmission loss of this circuit as a function of frequency is
shown on Fig. 2 together with the portion of the attenuation that is
contributed by conductance losses. Inasmuch as the intention is to
use these conductors at very high frequencies, a high grade insulating
material was used with the result that the conductance losses are small.
Fig. 1— Section of the New York-Philadelphia coaxial cable.
It should be noted that the attenuation increases very nearly as the
square root of the frequency.
In order to transmit high frequencies over long distances, a great
deal of amplification is obviously required. The New York-Phila-
delphia cable was initially equipped to handle a band of one million
cycles. Its overall attenuation at a million cycles is approximately
600 decibels. In order to reduce this to a usable amount 10 repeater
points were provided at intervals of about 10 miles each having an
amplification at the top frequency of about 60 decibels. These
repeaters were so designed that they provided less gain at low fre-
quencies than at the high frequencies, in approximately the same
degree as the line had less attenuation. To make up for certain
cumulative irregularities an equalizer was built and added to the
overall circuit. The net result was a transmission path which had
approximately zero loss over the whole band which it was desired to
use from 60 kc. to 1000 kc. as shown in Fig. 3.
440
BELL SYSTEM TECHNICAL JOURNAL
Another complicating factor is, however, involved, as is the case
with all long wire circuits, namely, the variation encountered with
changes in temperature. The loss in a 10-mile repeater section varies
materially from summer to winter. If the cable is hung overhead,
this variation is about as shown in Fig. 4 and amounts to a change of
about ± 7 per cent in the attenuation. If the line is buried under-
ground at the normal depth used for telephone cables, the actual
variation is about one-third as much. Automatic transmission regu-
lators were developed to compensate for these changes. These
200
400 600 800 1000 1200 1400
FREQUENCY IN KILOCYCLES PER SECOND
1800 2000
Fig. 2-
-Attenuation per mile of the New York-Philadelphia coaxial cable,
proportion of that attenuation due to conductance losses.
Also the
regulators depend, for their operation, on the transmission of a pilot
channel. At each point where it is desired to regulate the transmission,
the pilot channel is selected by a very narrow band filter and its
amplitude used to control an automatic device which changes the gain
of the repeater until the amplitude of the pilot at the output of the
repeater reaches a certain predetermined value. The regulators on
the New York-Philadelphia circuit have operated with such accuracy
that it has been unnecessary to make manual adjustments to take
care of temperature changes.
COAXIAL CABLE SYSTEAI FOR TELEVISION TRANSMISSION 441
The repeaters used along this route were novel in that most of them
were placed in small iron boxes located at convenient points along the
line. Power for their operation was transmitted at sixty cycles over
the coaxial cable itself. Figure 5 shows one such unattended repeater
located near Dunellen, New Jersey.
z
< +2.5
O
0
OVERALL, AFTER
EQUALIZATION
---K
■^
"-"^^
■^
[\
-2.5
-5.0
0
ION BA
1
-50
J -100
-n
J 150
-200
)
J -250
-300
-350
^
T^^
\^
95.5 MILES
LINE ALONE
\^
\
\
\
\
\,
>
\
y
-600
K
1
60
200 300 400 500
FREQUENCY IN KILOCYCLES PER SECOND
Fig. 3 — Overall attenuation of the New York-Philadelphia cable without repeaters
and the net attenuation after repeaters and equalizers had been provided.
The coaxial system between New York and Philadelphia was
designed to provide 240 telephone circuits. Skeleton terminal appa-
ratus was installed at New York and Philadelphia to test out such
operation. This apparatus has been described 2- 3. 4 Jn various papers
and Avill not be discussed here. Suffice it to say that methods and
442
BELL SYSTEM TECHNICAL JOURNAL
equipment have been developed which enable a wide band to be split
up so as to obtain hundreds of telephone channels.
For television transmission a quite different problem existed —
namely, can very wide band systems be used for the long distance
transmission of these complicated signals? In planning tests to be
significant of the operation of the cable system for such signals, it
was important to obtain as nearly as possible an ideal television
signal and as nearly as possible an ideal television receiver. In this
200 300 400 500 600 700 800
FREQUENCY IN KILOCYCLES PER SECOND
900 lOOOj 1100
1024
Fig. 4 — Attenuation of the New York-Philadelphia type of cable under widely
different temperature conditions.
way it was hoped that any defects in the cable transmission itself
could be discovered.
Signal Generator
Although television implies the transmission of an actual scene it
is much more satisfactory for engineering studies to transmit a motion
picture, since exactly the same picture can then be transmitted over
and over again as the circuit elements are changed or adjusted.
Moreover, it was decided to use mechanical scanning to obtain the
most nearly perfect signal possible, and with this form of scanning a
film can be much more brightly illuminated than an actual scene and
hence is much easier to use. Because of these factors a motion picture
film was chosen as the material for the recent experiments.
The scanning disk used in these tests was developed under the
direction of Dr. H. E. Ives at the Bell Telephone Laboratories. It
consists of a six-foot disk with a circle of 240 lenses near its outer edge.
COAXIAL CABLE SYSTEM FOR TELEVISION TRANSMISSION 443
The arrangenient is indicated schematically in Fig. 6. Light from a
powerful incandescent lamp behind the disk, passing through one lens
at a time, is focussed by the lens to form on the film a small dot of
Fig. 5 — Repeater near Dunellen, New Jersey.
light about three thousandths of an inch square. The lenses in the
disk are spaced by a distance equal to the width of the picture, or a
little less than an inch, and as the disk rotates, each spot is moved
rapidly across the picture. The film is carried at a uniform rate
444
BELL SYSTEM TECHNICAL JOURNAL
downward behind the disk at such a speed that the successive holes
throw their hght in successive rows across the picture one above
another. The film moves one frame for each revolution of the disk.
A photosensitive surface mounted behind the film picks up the light
transmitted through it, and produces a complex electric current
corresponding to the variations of light" in the picture. Figure 7 is a
photograph of the housing in which the disk is mounted. This
ii^----
PHOTO-SENSITIVE
SURFACE AND
ELECTRON
AMPLIFIER
Fig.'6 — Schematic diagram of the mechanical scanning arrangement used for
television testing.
scanning arrangement produced a picture of 240 lines, 24 frames per
second. It was recognized that 24 frames per second were not sufh-
cient to avoid flicker but this choice simplified the scanning apparatus
and it was believed would not interfere with engineering tests.
Signal Frequency Range
In order to understand what frequency is required to transmit an
image scanned in this way consider the diagram shown in Fig. 8.
COAXIAL CABLE SYSTEM FOR TELEVISION TRANSMISSION US
Of course, actual television will not ordinarily deal with such a picture,
but by means of it an approximate visualization of the problem can
be obtained. A desired definition of 240 lines was chosen and it was
decided that the requirement would be to transmit square picture
elements as indicated in the figure. The shape of the picture which
Fig. 7 — Photograph of the mechanical scanning disk and associated
experimental apparatus.
it was convenient to use with this scanning disk is wider than it is
high in the ratio of 7 : 6. This differs somewhat from the standard
aspect ratio of 4:3, but is easily taken into account. The total
7
number of picture elements in a frame is then 240- X 7 = 67200.
446
BELL SYSTEM TECHNICAL JOURNAL
If the smallest picture element to be transmitted is a single block
then the distribution of light and shade over the block is unimportant.
The average brightness over the block is what counts. Obviously,
a simple approximation is a sine wave as shown at the bottom of the
picture. This wave has 3^ cycle for each block and is as high a
frequency as there is any profit in transmitting for this diagram.
The top frequency needed for such a picture can then be calculated.
The number of square elements in the picture computed above is
240 X -s- ELEMENTS
TOP FREQUENCY - 240 X 240 X
■ X 24 =806.4 KILOCYCLES
Fig. 8 — Diagram illustrating the resolution of an image into picture elements and the
derivation of maximum frequency required for transmission.
67,200. As each of these elements represents 3^ cycle, this figure is
divided by 2, giving 33,600 cycles per frame. As similar frames are
reproduced 24 times a second, the result is 24 X 33,600 or 806,400
cycles per second. In a real moving picture, other frequency compo-
nents may exist at all other frequencies from 800 kc. down to and
including direct current. The direct current or zero frequency
component controls the general level of brightness of the picture.
Where the general level of brightness changes slowly, it results in a
component of very low frequency. A composite picture can be
COAXIAL CABLE SYSTEM FOR TELEVISION TRANSMISSION 447
imagined which will produce a pronounced component at any given
frequency, hence, it is deemed important to transmit the entire band
from 0 to 806 kc.
Receiving Device
At the receiving end an effort was made to obtain as high a degree
of fidehty of reproduction as possible. No small factor in the success
of the recent experiment was the special cathode ray tube, designed
by Dr. C. J. Davisson and used at the receiving end to display the
transmitted picture. Some of the features of this tube are indicated
schematically in Fig. 9. A stream of electrons from the cathode
passes through a series of electron lenses which focus a narrow beam
on an aperture .006" square. Between the lenses and the aperture,
however, are two modulating plates connected to the incoming circuit
in such a way that there appear on these plates potentials varying
according to the voltage of the incoming signals. The effect of
DEFLECTING
PLATES
MODULATING
PLATES
ELECTRON
LENS SYSTEM
Fig. 9 — Schematic diagram of the special cathode ray tube used for viewing
transmitted images.
potential on these plates is to deflect the electron beam, and the
conditions are such that at maximum strength of signal practically
the entire stream of electrons passes through the hole and forms a
brilliant spot of light on the front of the tube. As the signal decreases
in strength, the electron stream is more and more deflected; less
electrons pass through the aperture, and the illumination on the
sensitized end of the tube decreases.
In addition to these modulating plates, and placed between the
aperture and the front of the tube, are two other pairs of plates
mounted in planes at right angles to each other. The potential on
one of these sets of plates, controlled by a frequency of 5760 cycles,
which is the frequency at which successive lines are scanned, varies in
such a way that the beam of electrons passing through the aperture
is swept across the front of the tube from left to right, exactly in
synchronism with the scanning beam at the sending end. After the
beam reaches the farther side of the picture, the potential on the
448 BELL SYSTEM TECHNICAL JOURNAL
plates is suddenly changed, and the beam is rapidly moved back to
begin the next line. Due to a black mask down the far side of the
film being scanned, there is no signal during this very short period
while the voltage on the plates is changed, and thus the electron beam
is deflected from the aperture and is not visible on the front of the
tube during its return.
The potential on the other pair of plates is controlled at a frequency
of 24 cycles per second, which is the rate of scanning successive frames.
The effect of the potential on these plates is to deflect the electron
beam downward in synchronism with the motion of the film at the
sending end. This results in the passage of the electron beam across
the front of the tube in successive rows, one below another. After
the last row has been scanned, the voltage on the plates is changed and
returns to the value that causes the beam to appear at the top line
of the tube. A properly synchronized blanking-out pulse is introduced
between successive frames of the film, so that no signal is received
during this interval, and thus the passage of the electron beam from
the bottom to the top of the frame is not visible.
Figure 10 is a photograph of one of these cathode ray tubes. Due
to its superior design, the image is very sharp over the entire field and
a wide range of brightness is secured. The chief factors in its success
are the sharp focusing by the electron lenses, the linear deflection of
the beam at the aperture, and the great length of the tube, which
makes it necessary to deflect the electron beam over only a narrow
angle to cover the 7X8 inch field. Since this trial was a test to
determine the capabilities of the system, such matters as size and
cost, which would be important with commercial receivers, were not
controlling.
Modulation System
The frequency band which was generated at the sending end as
noted above was 0 to 806 kc. The coaxial cable system used could
not transmit this band, because repeaters were not designed to pass
frequencies below about 60 kc. This limitation was incorporated in
the original design because the cable offers insufficient shielding to
various disturbances at low frequencies. For television transmission
it was necessary, therefore, to raise the television signal band to a
higher frequency position before attempting transmission over the line.
A number of considerations led to the decision to raise the entire
frequency band 144 kc. for transmission over the coaxial cable.
Where such a wide frequency band is to be raised by an amount
less than the width of the band itself, a single modulation is not
generally satisfactory. The products of modulation include the
r
COAXIAL CABLE SYSTEM FOR TELEVISION TRANSMISSION 449
original frequency band as well as the upper and lower sidebands,
so that there will be a confusing jumble of frequencies in the modu-
lator output. For this reason a double modulation method was used
for the recent experiments.
Fig. 10— Photograph of one of the special cathode ray receiving tubes.
The modulating scheme employed can be followed with the help of
Fig. 11, which shows the two modulating steps at the sending end and
the two demodulating steps at the receiving end in four lines beginnmg
at the top. A carrier of 2376 kc. is used for the first modulation,
which results in a lower sideband from 1570 to 2376 kc. and an upper
450
BELL SYSTEM TECHNICAL JOURNAL
sideband from 2376 to 3182. The carrier itself is eliminated in the
balanced modulator. The output of this modulation is passed through
a filter, but because the two sidebands touch each other at 2376 kc,
the filter cannot cut ofif all the upper sideband. At the output of
this filter there is thus the lower sideband plus a small amount of the
lower part of the upper sideband. The upper sidebands from all
subsequent modulations are readily eliminated by filters because of
the wide separation.
TRANSMITTING END
RETAINED L»__ UPPER .^
\'^ SIDE-BAND ^
TELEVISION PASSED LOWER_^^_ J
mm
2376 (CARRIER)
(FIRST MODULATION)
REJECTED
[< UPPER >l
SIDE-BAND !
2520(CARRIER)
(SECOND MODULATION)
RECEIVING END
""^ 2376 (CARRIER)
(SECOND DEMODULATION)
FREQUENCY IN KILOCYCLES PER SECOND
Fig. 1 1 — Schematic diagram illustrating the processes of modulation and demodulation
used in transmitting television signals over the coaxial line.
The carrier for the second modulation is 2520 kc, and the lower
sideband extends from 950 down to 144 kc. plus a vestigial upper
sideband remaining from the first modulation which extends down to
120 kc. The high-pass filter following this modulation is accurately
designed to pass with controlled attenuation a group of frequencies
just above 144 kc. and the vestigial sideband. The resulting single
sideband, extending from 120 to 950 kc, is then passed over the
coaxial cable.
At Philadelphia the received signal, together with a carrier of
2520 kc, is applied to the first demodulator, and the lower sideband,
COAXIAL CABLE SYSTEM FOR TELEVISION TRANSMISSION 451
from 2400 down to 1570, is passed to the second demodulator where a
carrier of 2376 kc. is appHed. The lowest frequency of the lower
sideband, 1570 kc, is converted to 806 kc, becoming the highest
frequency of the final demodulated band. The frequencies from 2352
to 2400 kc. of the sideband before the second demodulation have been
attenuated somewhat by the high-pass filter following the second
modulator, and the second demodulating carrier, 2376 kc, falls in
the middle of this attenuated band as shown in inset No. 1. Fre-
quencies extending about 24 kc. above the carrier are inverted by the
demodulation, and superimposed upon the corresponding frequencies
just below the carrier. The magnitude and phase of these components
are proportioned by the high-pass filter and an equalizer so that the
overall result, when they are superimposed, is an essentially flat trans-
mission band from 0 to 806 kc.
The above steps of modulation involved a number of difficulties.
In the first place the signal level must be carefully controlled so that
on the one hand it does not sink into the background noise, while on
the other hand it must not be raised to such high levels that unwanted
modulation products are produced in too great magnitude. The first
modulator presents some special problems. It must accommodate all
frequencies from 0 to 806 kc. In order to eliminate the carrier, it
must be balanced to a very high degree — about 80 db in this case.
The reason the carrier must be so completely wiped out is that the 0
frequency component of the signal is identical with the carrier at the
output and hence the true d-c value of the signal must be exceptionally
free from carrier interference.
Referring to Fig. 12, the next piece of apparatus is a band filter to
eliminate the video signal and cut off the top edge of the band. Then
follows the 2nd modulator which is quite conventional. A low-pass
filter is next and is very important as it performs part of the function
of cutting off and adjusting the vestigial sideband. Then follows an
amplifier, a predistorting network to partially equalize the amplitudes
of the different components of the signal, an aperture equaHzer to
correct for the fact that the scanning spot is of finite size, a terminal
equalizer to make up for irregularities in the overall setup and other
amplifiers.
The carrier apparatus at the sending end is shown on Fig. 13.
It is mounted in rather conventional form except for the 1st modulator
which was arranged to minimize the effect of low-frequency vibrations.
At the receiving end about the same apparatus is required in the
inverse order and will not be discussed in detail.
452
BELL SYSTEM TECHNICAL JOURNAL
COAXIAL CABLE SYSTEM FOR TELEVISION TRANSMISSION 453
Carrier Supply
Another important feature of this system was the provision of
accurately spaced carriers for the various modulating operations. The
Bell System 4 kc. standard was used to produce a 72 kc. (18th har-
monic) base frequency signal which could be transmitted over the line
Fig. 13 — Photograph of the transmitting carrier television terminal with associated
sound apparatus.
and so tie the transmitting and receiving ends together. From this,
by harmonic generation, the carriers used in the two steps of modu-
lation were produced, being the 33rd and 35th harmonics.
To synchronize the scanning at the receiving end with the trans-
mitting disk required another direct tie. The disk is driven by a d-c.
454 BELL SYSTEM TECHNICAL JOURNAL
motor and its speed cannot be kept very constant. By means of an
auxiliary lamp and photocell, the frequency with which one scanning
line followed another — ^approximately 5760 per second — was obtained.
This was modulated with the 72 kc. mentioned above and transmitted
over the line as a lower sideband at 66.24 kc. At the receiving end
by demodulation, the exact line speed is obtained and used to drive
the horizontal sweep. The vertical sweep is obtained by generating
the 240th subharmonic of this — namely 24 c.p.s.
Line Equalization and Test Results
Returning now to the line transmission problem, the signals which
might be transmitted in the general case are indicated in Fig. 14.
They include the pilot channels used for automatic transmission
regulation at 60 kc. and 1024 kc. For convenience, one telephone
channel with a carrier at 64 kc. is indicated as an order wire, and a
PILOT FREQUENCY
I rSYNCHRONIZING CARRIER , PILOT
Ijr-CARRIER FREQUENCY
->{(*- ORDER-WIRE CHANNEL I
]W [<-PROGRAM CHANNEL T
III I k TELEVISION CHANNEL M
i
60|| , 84 120 FREQUENCY IN KILOCYCLES PER SECOND ^^° '^^^
6472
'I'
66.24
Fig. 14 — Frequency allocation for a television transmission system with associated
control circuits.
wide-band program channel with carrier at 84 kc. to transmit the
sound. These, of course, could be provided with ordinary telephone
facilities. The base frequency of 72 kc. and the disk synchronizing
sideband at 66.24 are also included. For the television signal itself
the band from 120 to 950 kc. is provided. Actually in the tests to
Philadelphia, automatic regulation was not needed and a separate
wire line was used for synchronization.
It was necessary to provide networks and equalizers to insure that
the coaxial line did not distort the ultimate image due to unequal
attenuation, resulting in amplitude distortion, or to unequal time of
transmission, causing phase distortion. The actual attenuation char-
acteristics of the line ^ and the overall result were shown above in
Fig. 3. The requirements for phase distortion are rather difficult to
meet. The details in the scanned picture result in various frequencies
of the electrical signal, and if these details are to appear in the repro-
duced picture in the same relative position as in the scanned picture,
it is essential that all frequencies be received in very closely the same
COAXIAL CABLE SYSTEM FOR TELEVISION TRANSMISSION 455
relative time relationship as they are generated. Referring back to
Fig. 8, it was assumed that no picture element could be displaced by
more than about half its width. This led to the decision to hold
frequencies between 806,000 and 3000 cycles to a delay distortion of
about 0.3 microsecond. For a similar degradation of detail in the
vertical direction, the permissible delay distortion is 280 times as
great which, in a system of this type, is very easily obtained. The
actual circuit roughly met these requirements as indicated by Fig. 15,
^+1
-!
-570
-565
-555
-550
-545
-540
-535
-530
-525
V 1 OVERALL, AFTER
V^.PHASE EQUALIZATION
1 ■
1
1
1
1
1
^
—
1
1
^
V
■>
1
1
LINE-REPEATERS 8.
LOSS EQUALIZERS
1
1
1
1
\
\
1
1
~- —
-
r
1
\
S
1
-TELEVISION BAND-
1
,1
V
1
1
\
1
1
1
\
S'
1
1
1
\
95.5 MILES
s^^^LINE ALONE
1
1
1
N
\
s.
1
1
1
s
V
V
1
TV
1
1
60
100 200 500 1000
FREQUENCY IN KILOCYCLES PER SECOND
Fig. 15 — Phase delay of New York-Philadelphia television circuit.
which shows the phase delay characteristics of the line, repeaters and
equalizers, and of the overall circuit at the frequencies used for trans-
mission.
Noise or interference is very annoying in television transmission;
and pattern, or single frequency interference, is particularly objection-
able. The permissible noise or interference depends on the amplitude
range of the reproduced picture. During these experiments, it was
found that a substantially linear response could be obtained over a
signal current range of about 20 db — a brightness ratio of 10 to 1.
456 BELL SYSTEM TECHNICAL JOURNAL
The actual reproduced pictures considerably exceeded this range; in
fact a brightness ratio of 50 or 100 to 1 was realized. In these tests
it was found desirable to hold random interference down about 40 db
below the maximum signal, and pattern interference down at least
15 db more.
Fig. 16 — Photograph of a two million cycle amplifier under development for
experiments on the coaxial cable.
The engineers who worked on the system, and outside experts who
observed it, expressed the opinion that the reproduced pictures in
Philadelphia were substantially the same as those seen on a similar
receiving device in New York, thus showing that the cable system
itself introduced no appreciable distortion.
COAXIAL CABLE SYSTEM FOR TELEVISION TRANSMISSION 457
Conclusion
As a result of the experimental transmission of the pictures over the
coaxial cable from New York to Philadelphia it has been proved that
wide-band signals of the type required for television can be satis-
factorily transmitted over a coaxial cable system, and that in such
transmission the distortion introduced by the wire line circuits can be
made so small as to be inappreciable, in its effect on the received
picture.
The work on these very wide-band systems has only begun and
repeaters and terminal apparatus are now under development capable
of handling wider bands of frequency. At the present time work is
under way on a two-million cycle system for telephone transmission
and a trial installation is being made between New York and Princeton.
The system will transmit a frequency band of about two million cycles
corresponding to a capacity of 480 telephone circuits. Repeaters on
this system will be about 5 miles apart and will consist of unattended
boxes somewhat smaller than the one-million cycle repeaters illustrated
above and placed either in manholes or on poles along the route.
Within these boxes there are placed two amplifiers, one for eastbound
transmission, the other for westbound, together with the necessary
filters and power supply apparatus. The actual amplifiers themselves
are quite small compact units one of which is shown in Fig. 16. Two
megacycles, of course, is not a sufficiently wide band to transmit the
present R.M.A. standard 441-line television signal, but is a logical
step toward more economical telephone circuits. Development is
also under way on amplifiers capable of transmitting three megacycle
bands of frequency, which should amply satisfy the requirements for
transmitting the 441-line television signals now envisioned as standard
by the television industry.
References
1. "Systems for Wide Band Transmission Over Coaxial Lines" by L. Espenschied
and M. E. Strieby, Electrical Engineering, Vol. 53, pp. 1371-1380, October
1934.
2. "A Million-Cycle Telephone System" by M. E. Strieby, Electrical Engineering,
Vol. 56, pp. 4-7, January 1937.
3. "A Carrier Telephone System for Toll Cables" by C. W. Green and E. I. Green,
Bell Sys. Tech. Jour., Vol. XVII, pp. 80-105, January 1938.
4. "Cable Carrier Telephone Terminals" by R. W. Chesnut, L. M. Ilgenfritz and
A. Kenner, Bell Sys. Tech. Jour., Vol. XVII, pp. 106-124, January 1938.
5. "Transmission Characteristics of the Coaxial Structure" by J. F. Wentz, Bell
Laboratories Record, Vol. XVI, pp. 196-200, February 1938.
Stabilized Feedback Oscillators
By G. H. STEVENSON
The author presents a mathematical consideration of the con-
ditions which insure constant frequency of the vacuum tube
oscillator under changes of electrode potentials or of the cathode
temperature. It has already been shown that the grid and plate
resistances may enter into the determination of the frequency.
The problem is treated here in the manner suggested in the recent
studies of feedback amplifiers. The conditions necessary for sta-
bility are developed in terms which are independent of particular
circuit configurations and are applicable to certain dissipative
circuits as well as to purely* reactive systems.
THE frequency deviations that accompany changes of the electrode
potentials or of the cathode temperature in many types of
vacuum tube oscillators have been recognized for some time as having
their origin in the variation of the internal resistances of the tube.
Llewellyn has shown ^ that both the grid and plate resistances may
enter into the determination of the frequency and, by treating the prob-
lem as one of network design, has demonstrated the possibility of mak-
ing the frequency substantially independent of the tube resistances.
He also devised a large number of oscillator circuits stabilized in this
way and established the conditions necessary for stabilization in each
case.
The problem is treated here in a somewhat more general manner sug-
gested by recent studies of feedback amplifiers.^ The conditions neces-
sary for stability are developed in terms which are independent of
particular circuit configurations and which permit their application to
certain types of dissipative circuits as well as to purely reactive systems.
While no new fundamental principles are presented, it is thought that
the restatement of the known principles in broader terms may be of
interest.
The mathematical theory will be developed for the case of the single-
tube oscillator circuit, since this is the form most generally used. The
extension of the theory to multiple stage circuits presents little or no
difificulty. The principal assumptions made are, first, that all of the
1 "Constant Frequency Oscillators," F. B. Llewellyn, Bell Sys. Tech. Jour.,
January 1932.
^"Regeneration Theory," H. Nyquist, Bell Sys. Tech. Jour., January 1932;
"Stabilized Feedback Amplifier," H. S. Black, Bell Sys. Tech. Jour., January 1934.
458
STABILIZED FEEDBACK OSCILLATORS
459
circuit elements except the tube resistances are linear and of lumped
character and, second, that modulation effects arising from the non-
linearity of the tube resistances may be neglected. The validity of the
second assumption is discussed in the appendix to the article by
Llewellyn noted above. Its use permits the treatment of the system
as though the resistances were actually linear but variable in magni-
tude in response to variations of the oscillation amplitude.
Theory
The essential features of the single-tube feedback oscillator are shown
in Fig. 1. The feedback network B is unrestricted in its configuration
E
\
2
O 1
1
2
FEEDBACK
NETWORK
(B)
Fig. 1 — Elements of a single tube feedback oscillator.
and complexity and may include the vacuum tube electrode capaci-
tances in addition to the external elements. The impedance system of
the tube is reduced to the plate and grid resistances R\ and R^ with uni-
lateral coupling between them, the latter being indicated by the inclu-
sion of a generator in series with the plate resistance. The voltage Ei
generated in the plate circuit is proportional to the voltage £2 between
the grid and the cathode and, when the system is oscillating, the latter
voltage is produced entirely by Ei as the result of the coupling through
the feedback network.
The condition for the existence of self-sustained oscillations is ex-
pressed very concisely by the familiar equation
M^ = 1,
(1)
wherein ^l and /3 denote the voltage transfer ratios in the vacuum tube
and in the feedback path respectively. The factor /^ is the negative of
the amplification constant of the tube, the negative sign taking account
of the phase reversal inherent in a simple triode. The transfer ratio /3
represents the ratio of £0 to Ei for transmission through the feedback
network.
460 BELL SYSTEM TECHNICAL JOURNAL
Since the transfer ratios ii and /S are both complex quantities, equa-
tion (1) expresses the two-fold requirement that the magnitude or
modulus of jx^ shall be unity and that its phase angle shall be zero.
Taking the factors separately, it follows that the modulus of ^ must be
the reciprocal of the modulus of ix and that the phase angles of the two
must be equal and of opposite sign. For the single-tube oscillator, the
phase angle of /3 must be 180 degrees since the phase angle of yu has
that value.
While the relationships stated above are of simple character, they
do not by themselves sufihce for the calculation of the oscillation fre-
quency from the constants of the tube and the external circuit. The
reason for this is that the values of the tube resistances Rx and R2 enter
into the determination of the frequency in the general case, and, since
these are dependent upon the oscillation amplitude, they cannot be
known until the final steady amplitude of the oscillations is known.
What happens in an actual oscillator circuit is that, as the oscillation
amplitude grows, after initiation, there is a mutual adjustment of fre-
quency and of the resistance values until a condition is reached under
which both requirements are met simultaneously. In the case of a
stabilized oscillator, since the frequency is independent of the tube
resistances, the conditions are simplified and the oscillation frequency
can be determined directly by means of the relationships stated above.
The non-linearity of the resistances afifects only the amplitude of the
oscillations.
The evaluation of ^ifi in terms of the impedance parameters of the
circuit permits the determination of the specific circuit conditions in
any case for the generation of steady oscillations. The determination
is simplified by the consideration that the factor ix has a constant
phase angle of 180 degrees so that the variation of the phase of /x(8 is
wholly that of the factor /3.
General Formulae for /x/3
The feedback path, or /3 circuit, is shown separately in Fig. 2, the
Fig. 2 — Simplified schematic of an oscillator feedback circuit.
notations being the same as in Fig. 1. The network B may be of any
degree of complexity, but may be assumed to be made up of lumped
STABILIZED FEEDBACK OSCILLATORS
461
impedances. In writing down the equations for the mesh currents, let
it be assumed that the circuit contains n meshes including the terminal
meshes and that the meshes are so chosen that the resistances R\ and
i^2 do not appear as mutual impedances. Designating the meshes in
which Ri and R2 appear as the first and second respectively, the mesh
current equations take the form
/l,
h,
Is-
• /„
Ri + Zn,
Z\i,
Zu ■
• Zi„
•2^21.
R'.
+ -2^22,
Z23 •
• Z2„
Z31,
Zz2,
Zs3 •
• Zsn
Z,,\,
Zn2,
Z„3 •
. 7
E
0
0
0
(2)
The subscripts of the Z's denote self and mutual impedances in ac-
cordance with the usual conventions, the latter being subject to the
reciprocal relationships characteristic of linear systems.
The solution of the above equations for the current I2. is
h =
- E^2l
A + RlR■,^n, 22 + i?iAn + i?2A22 '
(3)
where A is the determinant of the coefificients of equations (2) for zero
values of Ri and i?2, and the other determinants are the minors of A
obtained by crossing out the columns and rows indicated by the nu-
merical subscripts. Thus A21 is obtained by crossing out the second
column and the first row of A and An, 22 is obtained by crossing out the
first two columns and the first two rows.
Since, by definition,
^-~E''
equation (3) gives
— i?2A21
^ =
A -f i?li?2A„. 22 + Rl^U + R2A2i
(4)
The factor ^ is the negative of the amplification constant of the tube
and if the latter be denoted by a, the value of /i/3 becomes
i"/3 =
ai?2A2i
A + RiR^Au, 22 + RiAn + i?2A22
(5)
The determinants appearing in equations (4) and (5) can be ex-
panded by the ordinary processes to give expressions in terms of the
mesh impedances in any particular case. However, as they stand, they
462 BELL SYSTEM TECHNICAL JOURNAL
are significant parameters of the system and, for the present, need
no further expansion.
Another general formula for ju^S is obtained by making use of the
image parameters of the coupling network. If the image impedances
at terminals 1, 2, and terminals 3, 4, are denoted by Ki and K^,
respectively, and the image transfer constant by d, then
^^ ~ {K^K^ + R1R2) sinh e + {RiKi + R^K^) cosh d ' ^^
The two sets of parameters are related by the equations
All All.
22
^22 '-in, 22
tanh^0 = ^^^
A11A22
Equation (6) is useful in many cases because of the fact that the fre-
quency characteristics of the image parameters are well known for a
large number of circuit configurations, particularly those of wave
filters.
In dealing with many practical oscillator circuits, the simplifying
assumption may be made that the coupling network contains only pure
reactances. The determinants in equation (5) then become either real
quantities or pure imaginaries, thus making it easy to separate the real
and the imaginary parts of ^u/?. If the number of meshes in the jS cir-
cuit, or the number of rows or columns in the determinant A, is even,
then A will be real and if the number is odd A will be imaginary. The
determinant An, 22 will be of the same character as A, but will take
the opposite sign, and determinants An, A22, and A21 will be imaginary
when A is real and real when A is imaginary. Accordingly, equation
(5) may be transformed to
^^ {R^Du + R2D,,) + j{D - RrR^Du, 22) ' ^ ^
in which the D's are determinants of the mesh reactances correspond-
ing respectively to the A's of equation (5) having the same subscripts.
The phase angle of ^t/3, denoted by tp, is given by
D — RiRiDn, 22 /f.\
STABILIZED FEEDBACK OSCILLATORS 463
and the value of /Xj8, when tan ^ is zero, by
The angle - '2 25 A
^ ■ VV2
/
/
VS.y/2
/
J
X
FROM OBSERVATIONS WITH
/^^
DIFFRACTION APPARATUS.
/
o
SAME - PARTICULARLY RELIABLE
/Tr.
D
SAME -GRAZING BEAMS.
/xx x'^
jf X
®
FROM OBSERVATIONS WITH
REFLECTION APPARATUS
.05
/v
%
Fig. 4 — Test of the de Broglie formula X = hip = Ii/mv. Wave-length computed
from diffraction data plotted against l/F^'^, ( V, primar\' beam voltage). For precise
verification of the formula all points should fall on the line X = 12.25/ Vi/- plotted in
the diagram.
I would Hke also at this time to express my admiration of the late
Dr. H. D. Arnold, then Director of Research in the Bell Telephone
Laboratories, and of Dr. W. Wilson, my immediate superior, who were
sufficiently far-sighted to see in these researches a contribution to the
science of communication. Their vision was, in fact, accurate for
today in ours, as in other industrial laboratories, electron diffraction is
applied with great power and efficacy for discerning the structures of
materials.
But neither of this nor of the many beautiful and important re-
searches which have been made in electron diffraction in laboratories
in all parts of the world since 1927 will I speak today. I will take time
only to express my admiration of the beautiful experiments — differing
from ours in every respect — by which Thomson in far-away Aberdeen
also demonstrated electron diffraction and verified de Broglie's
formula at the same time as we in New York. And to mention, as
482 BELL SYSTEM TECHNICAL JOURNAL
closely related to the subject of this discourse, the difficult and beauti-
fully executed experiments by which Stern and Esterman in 1929
showed that atomic hydrogen also is diffracted in accordance with the
de Broglie-Schroedinger theory.
Important and timely as was the discovery of electron diffraction in
inspiring confidence in the physical reality of material waves, our
confidence in this regard would hardly be less today, one imagines,
were diffraction yet to be discovered, so great has been the success of
the mechanics built upon the conception of such waves in clarifying
the phenomena of atomic and subatomic physics.
Abstracts of Technical Articles from Bell System Sources
Stability of Two-Meter Waves} Charles R. Burrows, A. Decino
and LoYD E. Hunt. The continuous records of the field strength
received over a 60-kiIometer path on a frequency of 150 megacycles
for the year 1936 are analyzed. Preliminary comparison with other
paths of the same length indicate that the magnitude of the recorded
variations of the signals may be typical of paths of this length.
A reduction in the path length by a factor of two reduced the
fading range in decibels by a factor of five.
The results are found to be in agreement with an earlier formula.
Fading reduced the field 7 decibels below the average value 1 per cent
of the time.
Loudness, Masking and Their Relation to the Hearing Process and
the Problem of Noise Measurement} Harvey Fletcher. It is shown
in this paper how to define loudness and loudness level in a quantitative
way. Definite procedures are given for determining experimentally
the loudness level of any sound heard by any person. For a typical
observer a true loudness scale is developed. The relation of the scale
to the loudness level scale is determined experimentally. The scale
has been found to be very useful for calculating loudness from the
noise spectrogram, the noise audiogram, or the overtone structure of
the sound.
The relation between the masking and the loudness produced by
a sound has been quantitatively determined and a formula deduced
from this relation which has proved useful for calculating the loud-
ness. This formula may be applied with equal success to a normal
ear and also to a deafened ear. Evidence has been given that the
masking expressed in decibels produced upon any pure tone is equal
directly to the agitation of 1.1 per cent of the total nerve endings
expressed in decibels above the threshold value for such a patch and
at the position where such a tone would be sensed. These loudness
relations throw light upon some of the important processes involved
in hearing. In particular the data from the masking effects of thermal
noise were used to calculate the relation between the position of
1 Proc. I. R. E., May 1938.
2 Jour. Acous. Soc. Amer., April 1938.
483
484 BELL SYSTEM TECHNICAL JOURNAL
maximum stimulation on the basilar membrane and the frequency
of the tone producing the stimulation.
Pick-up for Sound Motion Pictures {Including Stereophonic) .'^ J. P.
Maxfield, a. W. Colledge and R. T. Friebus. Although the basic
principles underlying sound pick-up for motion pictures have been
understood for some time, the ability to carry them out completely
in the presence of the requirements of artistry, photography, lighting,
etc., has constituted a difficult problem. The paper discusses some
of these problems, particularly with respect to the acoustics of produc-
tion sets and scoring stages. The problems of stereophonic repro-
duction are also discussed in some detail.
Practical Application of Telephone Repeaters and Carrier Telephone
Systems} J. A. Parrott. The paper discusses engineering problems
in the application of telephone repeaters and carrier systems with
which railroad communication engineers recently have been particu-
larly concerned. The first part of the paper deals with crosstalk,
noise, balance and overloading considerations in the design of re-
peatered circuits, particularly from the standpoint of selecting the
locations of repeaters to obtain the most satisfactory results on existing
lines. The importance of securing test data on the wire facilities to
aid in this design work as well as to serve as a guide in improving
circuit conditions is emphasized.
The second part of the paper briefly discusses the application of
the HI carrier telephone system and provides transmission data for
the preliminary design of the layout of such systems. The Type D
and KIO carrier transpositions are described and features of particular
interest in their possible use on railroad facilities are discussed.
Sorption of Water by Rubber} R. L. Taylor and A. R. Kemp,
The effect of several variables on the rate of sorption of water by
rubber is discussed. Expressions based on short-time immersion tests
are derived which permit calculation of the water content after an
extended period of immersion under fixed conditions of temperature
and vapor pressure. A sorption coefficient by which one material
may be compared with another is suggested, and its application to
practical problems is considered.
3 Jour. S. M. P. E., June 1938.
* Proc. Assoc, of Amer. Railroads, Telegraph and Telephone Sec., October 1937.
^ Indus. & Engg. Chemistry, April 1938.
ABSTRACTS OF TECHNICAL ARTICLES 485
Chemical Studies of Wood Preservation — The Wood-Block Method of
Toxicity Assay.^ Robert E. Waterman, John Leutritz and Caleb
M. Hill. Actual decay resistance of treated wood is used as the
basis for a simple laboratory technic in the assay of materials advo-
cated for the protection of wood. In its present stage of development
the test is a valuable tool in wood preservation studies.
^ Indus. &• Engg. Chem., Anal. Ed., June 15, 1938.
Contributors to this Issue
Julian Blanchard, A.B., Trinity College (now Duke University),
1905; A.M., Columbia University, 1909; Ph.D., 1917. Professor
of Engineering, Trinity College, 1909-1912; Research Assistant in
Physics, Columbia University, 1912-1915. Physicist, Research Lab-
oratory, Eastman Kodak Company, 1915-1917; Engineering Depart-
ment, Western Electric Company, 1917-1925; Bell Telephone Labora-
tories, 1925-. Dr. Blanchard's work has been concerned primarily
with special studies in connection with the development of vacuum
tubes and radio.
B. L. Clarke, B.S., George Washington University, 1921; M.A.,
Columbia University, 1923; Ph.D., Columbia University, 1924. Bell
Telephone Laboratories, 192 7-. Dr. Clarke has been in charge of the
work in analytical chemistry since 1930.
C. J. Davisson, B.Sc, University of Chicago, 1908; Ph.D., Prince-
ton University, 1911; Instructor in Physics, Carnegie Institute of
Technology, 1911-17. Engineering Department of the Western Elec-
tric Company, 1917-25; Bell Telephone Laboratories, 1925-. As
Research Physicist, Dr. Davisson is engaged in work relating largely
to thermionics and electronic physics.
In 1928 the National Academy of Sciences awarded the Comstock
Prize to Dr. Davisson "for the most important discovery of or investi-
gation in electricity or magnetism or radiant energy" made in this
country during the preceding five years, for his work in this field.
In 1931 he and Dr. L. H. Germer received the Elliott Cresson Medals
from the Franklin Institute, Philadelphia, and in 1935 he received
the Hughes Medal of the Royal Society of London.
W. G. GusTAFSON, B.S. in Electrical Engineering, Union College,
1927; Columbia University, 1929-36. Bell Telephone Laboratories,
1927-. Mr. Gustafson is engaged in work relating to the development
of transformers and repeating coils for communication purposes.
W. Herriott was engaged in astronomical research at the Allegheny
Observatory from 1914 to 1917. Research in astronomical and aerial
photography at the Research Laboratories of the Eastman Kodak
Company followed from 1917 to 1920. Between 1920 and 1925 he
486
CONTRIBUTORS TO THIS ISSUE 487
was engaged in the development of military instruments and of optical
apparatus for microscopy, photography and motion pictures at the
Bausch and Lomb Optical Company. During the following three
years he was in charge of the Scientific Department of the Fairchild
Aerial Camera Corporation. In 1928 he joined the engineering de-
partment of Electrical Research Products, Inc., coming to the Bell
Telephone Laboratories in 1929 to work on optical and photographic
problems associated with sound picture apparatus development. In
October 1936 he transferred to the Materials Group of the Electro-
mechanical Division of the Telephone Apparatus Development De-
partment.
A. H. Inglis, B.A., Yale University, 1914. Western Electric Com-
pany, Engineering Department, 1914-17. Signal Corps, A.E.F.,
1917-19. American Telephone and Telegraph Company, Department
of Development and Research, 1919-34; Bell Telephone Laboratories,
1934-. Mr. Inglis has been concerned with both equipment and
transmission matters of station apparatus, latterly as Station Instru-
mentalities Engineer.
W. C. Jones, B.S. in Electrical Engineering, Colorado College,
1913. Western Electric Company, Engineering Department, 1913-
25 ; Bell Telephone Laboratories, 1925-. As Transmission Instru-
ments Director, Mr. Jones is concerned with the development of
telephone instruments and similar devices.
H. C. Montgomery, A.B., University of Southern California, 1929;
M.A., Columbia University, 1933. Bell Telephone Laboratories,
1929-. Engaged at first in studies of hearing acuity and related
problems in physiological acoustics, Mr. Montgomery has been occu-
pied more recently with the study and analysis of speech.
A. E. RuEHLE, B.S., University of Idaho, 1930. Bell Telephone
Laboratories, 1930-. Mr. Ruehle's work has been chiefly concerned
with applications of the methods of physical chemistry to chemical
analysis.
G. H. Stevenson, B.Sc. in Engineering, University of Glasgow,
Scotland, 1906; Instructor in Electrical Engineering, University of
Glasgow, 1906-07. Messrs. Barr and Stroud, Glasgow, 1907-11.
Western Electric Company, Engineering Department, 1911-24; Patent
Department, 1924-25. Bell Telephone Laboratories, Patent Depart-
ment, 1925-. Mr. Stevenson's work has to do with patent matters
488 BELL SYSTEM TECHNICAL JOURNAL
in the fields of wave transmission networks and radio transmission
systems.
M. E. Strieby, A.B., Colorado College, 1914; B.S., Harvard, 1916;
B.S. in E.E., Massachusetts Institute of Technology, 1916; New York
Telephone Company, Engineering Department, 1916-17; Captain,
Signal Corps, U. S. Army, A. E. F., 1917-19. American Telephone
and Telegraph Company, Department of Development and Research,
1919-29; Bell Telephone Laboratories, 1929-, Mr. Strieby has been
associated with various phases of transmission work, more particularly
with the development of long toll circuits. At the present time, in his
capacity as High Frequency Transmission Engineer, he directs studies
of new and improved methods of carrier frequency transmission over
existing or new facilities.
The Bell System Technical Journal
Vol. XVII October, 1938 No. 4
Ultra-Short-Wave Transmission and Atmospheric
Irregularities
BY C. R. ENGLUND, A. B. CRAWFORD AND W. W. MUMFORD
Results of an ultra-short-wave fading study are here reported.
Transmission was carried out in the range of 1.6 to 5.0 meters, over
a 70 mile (112.6 kilometer) ocean path, on 106 days during a period
of two years. Both horizontal and vertical polarizations were used
and during part of the time a 6-megacycle amplitude, 120-cycle,
frequency modulated transmission was added, for the cathode-ray
tube observation of the frequency characteristics of the radio path.
On 45 mornings records were taken, on vertically polarized radia-
tions, during the flight period of the Mitchel Field Weather Bureau
plane.
Fading was found present practically all of the time. Amplitude
changes up to 40 db and fading rates up to 5 fades per minute were
found. Simultaneous transmission of the same wave in two polar-
izations, and of two waves of different wave-length in the same
polarization showed that the horizontally polarized component was
practically always, and the shorter wave-length one was usually
the worse fader of the pair. The greater part of the time there
was no correlation between the fading of these radiation pairs;
occasionally, however, and for the slow, smooth amplitude,
undulating type of fading, coincidence was observed. The fre-
quency sweep patterns showed multiple signal components to be
present, with various degrees of relative phase retardation.
A tentative explanation is proposed for these phenomena. This
theory assumes the presence of a refracted-diffracted signal com-
ponent, transmitted along the earth's surface and calculable in the
manner of Wwedensky, Van der Pol and Gray, and one or more
signal components reflected from air mass boundaries. The air-
plane results are shown to be in reasonable agreement with the
frequency sweep observations. Boundary heights from 5.5
kilometers down to 1.9 kilometers are measured; below 1.9 kilo-
meters other boundaries are indicated. The receiver band, flat over
two megacycles, sets the low height limit of resolution of reflecting
boundaries at 1.9 kilometers. Most of the boundaries are at the
lower heights.
489
490 BELL SYSTEM TECHNICAL JOURNAL
A discussion is given of some observations of signal fading at
various wave-lengths which have been reported by other ob-
servers, and which are apparently referable to the same mechanism
as is here proposed.
Introduction
IN an earlier paper ^ experimental data were presented which indi-
cated that the transmission of ultra-short-wave signals was de-
pendent upon the state of the atmosphere, in particular upon its water
vapor content. The present paper contains the results of a continua-
tion of this work where a two-year survey of ultra-short-wave trans-
mission over a 70-mile (112.6 km.) ocean path was carried out. Trans-
mission was had on 106 days during this period.
In planning this work, preparation was made for seeking a correla-
tion between atmospheric structure and signal intensity; but from the
very first transmission fading was found, and this fading was so per-
sistent and intense that the work became essentially a fading study.
In the following paragraphs there are discussed, in the order named,
Antennas and Locations; Apparatus and Operation; General Charac-
teristics of Fading, with samples of records taken; Polarization Effect
on Fading, with sample records; Wave-length Effect on Fading, also
with records; Distance and Antenna Height Effects on Fading;
Frequency Sweep Patterns of Fading, with sample records; and the
logs taken during the flights of the U. S. Weather Bureau airplane for
taking free air data. The presentation of experimental data is then
interrupted to present a theory which explains several of the experi-
mental observations. This is followed by further experimental results
and checks, and concluding remarks.
Antennas and Locations
Figure 1 shows the layout of the radio circuit. The transmitter
was erected at Highlands, New Jersey on the edge of a steep hillside.
This edge made an angle of about 45° with the transmitter-receiver
direction. Below the edge of the hill lay a strip of land slightly above
sea level (seven to eight feet) and beyond was Sandy Hook Bay.
The altitude at the antenna foot was 119 feet. The antennas con-
sisted of a vertical rhombic terminated in its surge impedance with
carbon lamps, a horizontal rhombic with the same termination, an
unterminated inverted "Vee" antenna and a half- wave doublet.
This doublet was equipped with a flexible transmission line which
permitted it to be raised to the top of the antenna supporting mast.
These antennas were supported on a central 60-foot (18.3 meter) lattice
mast surrounded by four 30-foot (9.15 meter) poles.
I
UL TRA-SHOR T-WA VE TRA NS MISSION
491
The receiver was located on a plot of land at East Moriches, Long
Island, New York. This plot was immediately at the edge of Moriches
Bay and was only slightly (approximately four feet) above sea level.
The same antenna equipment was supplied here as at the transmitter.
Except for the transits across Sandy Hook, Fire Island Beach and
Smith Point, the wave path was over sea water. A second receiving
site at West Sayville, at the edge of Great South Bay, was briefly
occupied, using portable receiving equipment. This site was 52^
miles (85 km.) from Highlands.
Fig. 1 — Map of ultra-short-wave transmission path between Highlands, New Jersej',
and East Moriches, Long Island.
Apparatus and Operation
In all, three transmitters were installed at Highlands. The first
one, of 100 watts output, covered the wave-length range of 5.0 to 3.5
meters. It was equipped with a motor-driven single-turn short-
circuit loop which, coupled with the tank circuit coil, produced a 120-
cycle frequency modulation of six megacycles amplitude. For cali-
bration purposes there was added a low-gain double-detection receiver
which used an intermediate frequency of one megacycle and was con-
nected so as to pick up an input from the transmitter. The beating
oscillator of the receiver was set for the center of the transmitter fre-
quency sweep and the receiver output triggered a gas tube connected
492 BELL SYSTEM TECHNICAL JOURNAL
to the transmitter tube grids. The transmitter grids thus received a
voltage pulse each time that the transmitter frequency passed through
one megacycle above or below the beating oscillator frequency. Each
transmitter frequency sweep was thus marked with two pulses spaced
two megacycles apart.
The second transmitter had Lecher wire tuning elements, covered
the wave-length range of 3.5 to 1.2 meters and had a power output of
30 watts at 1.5 meters. It was in operation simultaneously with the
first transmitter for six months and then was replaced by transmitter
No. 3.
The third transmitter was coil tuned, covered the wave-length range
of 4.9 to 2.8 meters and had a power output over this range of 55 watts
down to 35 watts. It was operated simultaneously with the first
transmitter except for the first six months.
All three transmitters were arranged for voice modulation through a
simple grid input, and the first one was thus used for one-way com-
munication during the entire period of operation.
Normally, unmodulated waves were transmitted and were observed
as rectified direct current in the output of the double detection re-
ceivers. These receivers had attenuators, variable in steps of 1 db,
in the intermediate frequency amplifier circuits and the attenuators
were geared to the pens of manual recorders. The operators kept the
output current constant by means of the attenuators just mentioned,
and there resulted a record of signal amplitude versus time. Some use
was made of the Esterline-Angus type of milliampere recorder for
automatic recording but no linear scale recorder of this type could
handle the amplitude range of the fading encountered.
For the reception of the frequency modulated transmission a tuned
radio-frequency receiver, with a three-megacycle band-width centered
on 66 megacycles (4.55 meters), was constructed and its rectified output
was applied to one pair of plates of a cathode ray oscillograph. A
linear sweep voltage, manually synchronized with the transmitter
60-cycle power voltage, was applied to the second pair of plates. The
oscillograph pattern thus pictured the frequency-amplitude charac-
teristic of the radio circuit in toto. Over the frequency range where
the receiver band was flat (two megacycles) the curve gave the ap-
parent ether characteristic. With a motion picture camera this
characteristic was permanently recorded.
Fading Characteristics, General
The fading was always slow compared with that observed on short
waves. Except for the rapid fluctuations produced by airplane reflec-
UL TRA -SHOR T- WA VE TRA NSMISSION
493
tions, a record speed of ^ inch (1.6 cm.) per minute was sufficient.
This was our standard speed. Amplitude changes up to 40 db and
fading rates up to 5 fades per minute were observed.
It is difficult to describe the fading in any other way than by the
records. From a transmission standpoint a curve giving the per cent
of time during which the signal is above the abscissa value is useful.
a
^^
^^
4.7 METERS
JULY 23, 1934
5=00 A.M.
EASTERN STANDARD TIME
Fig. 2 — Fading extremes, vertically polarized transmission; inverted "V" antennas.
8:00 A.M.
e;30 A.M.
EASTERN STANDARD TIME
Fig. 3 — Extreme amplitude, normal fading rate, vertically polarized transmission;
inverted "V" antennas.
494
BELL SYSTEM TECHNICAL JOURNAL
>
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11-30 A.M.
EASTERN STANDARD TIME
Fig. 4 — Development of "scintillation" fading on vertically polarized
ULTRA-SHORT-WAVE TRANSMISSION
495
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AUG. 29, 1934
3:30 P.M.
EASTERN STANDARD TIME
transmission, 4.74 meters wave-length; inverted "V" antennas.
496
BELL SYSTEM TECHNICAL JOURNAL
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SEPT. 2
1^1934
40
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4:30AM
5:00AM 5 30AM
EASTERN STANDARD TIME
Fig. 5 — Twenty-four hour run, vertically polarized
ULTRA-SHORT-WAVE TRANSMISSION
497
,— .
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1,1934
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4:30PM
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5:30PM 6:00PM
EASTERN STANDARD TIME
6:30PM
7:00PM
transmission, 4.74 meters wave-length.
498 BELL SYSTEM TECHNICAL JOURNAL
Such a curve can also serve to check on the theoretical explanation of
the cause of fading in certain cases. Thus if the fading is due to the
combination of two radiation components in assigned random ampli-
tude relation and arbitrary or random phase relation, a curve can be
calculated from probability considerations and compared with the
experimental curve. ^ Such a simple mechanism was inadequate for
our fading most of the time. Moreover, the fading changed enor-
mously from day to day. It is hoped that the samples given in the
figures will give an adequate idea of this phenomenon.
Only rarely was fading practically absent for periods of an hour or
two. Such a period is illustrated in Curve a, Fig. 2. Two days later
the extreme fading of Curve h was recorded. It is significant, as will
later appear, that the non-fading situation was the one of higher signal.
The amplitude range of curve h is nearly normal; the fading rate is
much greater than normal, for vertical polarization. In Fig. 3 the
fading rate is normal but the amplitude range is excessive. In Fig. 4
a characteristic type of fading, which we have termed "scintillation,"
is recorded. In this case the fading, initially erratic and of a fairly
wide amplitude range, subsides in a characteristic manner to a steady,
fast rate oscillation, or scintillation, of moderate amplitude.
In Fig. 5 a 24-hour run is recorded. The rambling erratic character
of the fading is well shown here. Characteristic deep short-period
minima occur at intervals, occasionally they are twinned, some of them
have a fine structure at the bottom. There are several "dropouts"
where the signal practically disappeared.
No sunrise-sunset variations in fading were noticed, though looked
for. Diurnal variations could not be established since automatic
recording was not available. A seasonal falling off in average signal
was noticed in the winter; the 1.6 meter wave, because of its normally
low level, dropped below the noise level in the winter of '34-'35. No
effect of ocean waves, clouds, or other visible weather phenomena
could be established. It is true, however, that to be certain of the
non-effect of such phenomena as clouds, a cloud observer at the mid-
way point should have been present. In so far as cloud layers make
air mass boundaries visible they may well affect the transmission.
Cloud bottoms which represent merely the adiabatic dew point level
should apparently not cause much signal reflection at these wave-
lengths.
Effect of Polarization on Fading
After some preliminary experimenting it was found that comparisons
of two transmissions were worthless unless made on simultaneous
recordings. The recorders were therefore fitted with telechron motors
ULTRA-SHORT-WAVE TRANSMISSION
499
operating on a circuit of the Patchogue division of the Long Island
60-cycle power network. The resulting timing was faultless and by
transmitting the same radiation on crossed antennas, and receiving the
vertical and horizontal components separately, a comparison was
obtained.
In general the horizontal component showed the worse fading, more
fades per minute and greater amplitude range. This was always true
when the fading on vertical polarization was bad. There was then no
noticeable coincidence between the two. When the fading had a
smooth long period fade, or "roller," superposed on a short period
oscillation, or "line structure," there was at times coincidence between
the roller components. Occasionally, with fine structure absent and
I
9:30PM lOOOPM
Fig. 6 — Composite bad fading, horizontally polarized transmission.
500
BELL SYSTEM TECHNICAL JOURNAL
aa Ni Hi0N3aj.9 ivnois 3aiivi3h
ULTRA-SHORT-WAVE TRANSMISSION
501
moderate roller fading, a good coincidence between the two records
resulted. This is discussed later.
Figure 6 is a sample of fading on horizontal polarization, at its worst.
This particular specimen shows the superposition of roller and fine
structure fading very well. No vertical-polarization record was taken
along with this. Figure 7 shows a typical example of fading simul-
taneously observed on vertical and horizontal polarization during bad
fading conditions. There is no coincidence. Figure 8, on the other
hand, records an unusual condition when a mild roller type of fading
shows a good coincidence on two polarizations.
30
20
X 10
^\/A
/X-~s
^^r-
-^
^~\
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VERTICAL
3.5 METERS
\l
SEPT 26, 1935
50
> 40
30
20
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x/"
^
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-J
HORIZONTAL
3.5 METERS
-^v
SEPT 26, 19
35
1 = 30 RM.
2:00 RM.
EASTERN STANDARD TIME
2:30 RM.
Fig
. 8 — Comparison of simultaneous mild roller fading on horizontally and vertically
polarized transmissions.
Effect of Wave-length on Fading
The double wave-length records are not as contrasty as the double
polarization ones. In general the shorter wave has the worse fading,
either as higher fading rate, greater amplitude oscillation or both, and
the greater the wave spacing the more certain this is to be true. Ex-
ceptions have occurred, however, where the fading was much the same,
and one record was obtained where the fading rate on 4.7 meters was
noticeably greater than on 4.5 meters.
Our first simultaneous records were taken at a wave-length ratio of
3 to 1 (4.7 to 1.58 meters) where the fading on the shorter wave was
502
BELL SYSTEM TECHNICAL JOURNAL
always worse. The remaining observations were confined to wave-
length ratios of 1.5 to 1 and less. A comprehensive set of records was
obtained for moderate to small wave-length spacings, down to 1 per
cent difference. These records are all for vertical polarization. The
few records taken on horizontal polarization happened to be obtained
when the horizontal fading was much worse than the vertical fading
and the records are too rough for good comparisons.
For these small wave-length-difference records the types of fading
are more likely than not to be similar on the two wave-lengths. That
is, the fading rate and amplitude excursion will average up much the
same. More rarely, there will be a similarity between the two fading
tracks which is evident to the eye, sometimes as a "retarded" simi-
6=30 A.M.
EASTERN STANDARD TIME
7:00 A.M.
Fig. 9— Comparison of simultaneous fading on two well spaced wave-lengths,
vertically polarized transmission.
larity. Occasionally, and usually on the roller type of fading, there
will be a marked coincidence between the two records; this coincidence
will be better the milder the fading and the smaller the wave-length
spacing. Genuine identity was never recorded on different wave-
lengths even down to 1 per cent difiference. With scintillation, coin-
cidence was difficult to demonstrate; a similarity on the major swings
was all that was shown.
Figure 9 shows a very marked difference between 4.7 and 3.0 meter
fading. This is one of our most contrasty records. Figure 10 shows
very slow fading, on two occasions, with wave-length differences of
ULTRA-SHOR T-WA VE TRA NSMISSION
503
approximately 1 and 4 per cent respectively. There is good coinci-
dence. Figure 11 shows active fading on short rollers for 4.7 and 4.65
meters, a wave-length difiference of approximately 1.1 per cent. There
is agreement in major features. Figure 12 shows a case of scintillation
2:00 RM.
2:30 RM
3:00 RM.
< 40
O
30
20
10
^
./^^
/4..7 METERS
"V
^
V
V
\
V
V
4.5 METERS
A
\a
/
AUG. 8, 1935
V
5:00 A.M.
5:30 A.M.
EASTERN STANDARD TIME
6:00 A.M.
Fig. 10 — Comparison of simultaneous slow fading on two slightly different wave-
lengths, vertically polarized transmission.
superposed on mild rollers, again for 4.7 and 4.65 meters. The time
scale is here magnified three times. An in and out similarity can be
seen, especially for the rollers. In the section on theory these simi-
larities are further discussed.
504
BELL SYSTEM TECHNICAL JOURNAL
30
20
4.7 METERS
10
^V\A/VyVM|
4:00 P.M
4=30 P.M.
EASTERN STANDARD TIME
5:00 P.M.
Fig. 11 — Comparison of simultaneous active fading on two slightly different wave-
lengths, vertically polarized transmission.
30
20
10
4.7 METERS
f^jV,
'^"^^Vv./f
ry
0f^
yf"^
ifVv
1
1
1
MAR.
6. 1935
1:30 P.M.
1:35 RM. 1:40 RM.
EASTERN STANDARD TIME
1:45 RM.
Fig. 12 — -Comparison of simultaneous "scintillation" fading on two slightly
different wave-lengths, vertically polarized transmission. The time scale has been
expanded.
Effect of Distance and Antenna Height on Fading
In planning this work a survey for a receiving site was made by
means of a portable receiver in a car. Later, simultaneous reception
ULTRA-SHORT-WAVE TRANSMISSION 505
was had at East Moriches and West Sayville, on three days. The
survey data were not sufficient to establish any proposition beyond
the statement that the signal strength fell rapidly with distance, with
the intensity of fading coming up as the signal fell. The simultaneous
two-distance recording showed random fading between the two
records with less fading amplitude at the shorter distance. The
fading rate was about the same. Unfortunately the recording took
place under scintillation conditions, thus giving very poor records for
comparison purposes.
By mounting two linear doublets on the 60-foot lattice mast simul-
taneous recording at two heights was carried out. For the two dou-
blets (horizontal, at 14 and 52 feet respectively), a signal level difference
of 12 db was observed, in favor of the higher antenna. The fading on
the two records was identical. It may be added that, on calibrating
the car receiver at East Moriches before moving to West Sayville,
identical fading records were obtained with the two antenna systems
150 feet (45.7 meters) apart and substantially broadside to the
radiation.
Frequency Sweep Patterns of Fading
The frequency sweep patterns were of many types, from slow to fast
fading and from shallow to deep fading. Apparent path differences
from 600 meters down to a few meters occurred. The patterns were
usually complicated, indicating that more than two components were
present. There is no reason to believe, however, that they were not
all due to wave interference.^*
On three days the predominant pattern was simple enough to be
referable to two waves with a path difference consistently greater than
75 meters. These will be referred to later. In Figure 13 are given
three sample runs illustrating a two-component pattern, a three-
component pattern with two of the components forming a small path
difference pair, and a multiple component pattern. The receiver
characteristic is dotted in on one curve of each set.
Logs During Weather Bureau Airplane Flights
On forty-five mornings recording was carried out during the period
of flight of the Mitchel Field Weather Bureau plane. This plane takes
off about dawn every morning, when flying is possible, and by means
of a meteorograph obtains records of air pressure, temperature and
humidity, up to an altitude of about five kilometers. A record of the
fading, on 4.7 meters and vertical polarization, was obtained for each
of these mornings. In addition, on twenty-six mornings frequency
506
BELL SYSTEM TECHNICAL JOURNAL
sweep patterns were photographed at or shortly after the time of
flight. These sweep patterns were all on horizontal polarization.
From the meteorograph data, kindly furnished us by the United
States Weather Bureau, the dielectric constant of the air has been
calculated ^ and plotted as a function of the altitude. On twenty-four
days there were, above an altitude of 400 meters, changes in the
dielectric constant curves equivalent to discontinuities of Ae ^ 10~^.
Heights up to 3200 meters were recorded for these. Typical curves
JUNE 27,1936
2 COMPONENT
JUNE 28jf936
3 COMPONENT
603:55
64 66 69
MAY 19,1936
MULTI-COMPONENT
64 66 69
FREQUENCY IN MEGACYCLES
64 66 69
Fig. 13 — Three sequences of frequency sweep patterns. Horizontally polarized
transmission, 4.55 meters mean wave-length.
are given in Fig. 18 on the left-hand side. On four days there were
small boundaries with Ae < 10~^; on two days there were possible but
not definite boundaries, the experimental points being too widely
separated in altitude for precision; on five days there were possible
boundaries below 400 meters altitude and on ten days the refractive
ULTRA-SHORT-WAVE TRANSMISSION 507
index-height relation was an approximate exponential one without any
evident boundaries. These data will be referred to later.
Theory
The fading phenomenon was explicable in several ways. In our
previously cited work ^ we found that variable atmospheric refraction
was present, the airplane carried receiver being up where the refracted-
diffracted field strength was high and dominant. In general variable
refraction would be expected to be a slow phenomenon, operating in
hours, or even days, rather than in minutes, and much too slow to
explain five-cycle-per-minute oscillations, for example.
Another explanation was air-mass boundary reflection (or refrac-
tion),^ such a boundary readily explaining the rate of signal variation.
No Kennelley-Heaviside layer reflection was in question ; this had been
quickly ruled out by the experimental data. When, therefore, we
elected to transmit the frequency modulated signal, already described,
and the oscillograph revealed a cyclic maximum-minimum frequency
characteristic of the other path itself, it was evident that there was no
possibility other than wave interference left — interference presumably
between a direct-diffracted and one or more boundary-reflected
components.
These boundaries have apparently not been positively identified at
longer wave-lengths and for that reason we have tried to get some
further experimental contact with them. Attempts, since the closing
down of the Atlantic Highlands-East Moriches circuit, to demon-
strate an air-mass boundary, any boundary whatever, by high-angle
transmission, have failed. No reflected components have appeared.
Of course an illy defined, or diffuse, boundary will operate in this
manner since only for near grazing incidence can such a boundary give
the appearance of a discontinuity for the incident radiation.
If we assume such a boundary a few kilometers up, and assign to it a
relatively small discontinuity in index of refraction, compared with
that of an earth or sea water boundary, then the four components of
Fig. 14 will be the only important boundary reflected ones for a radio
circuit such as ours. We now, fortunately, have theoretical formulae *• ^
for computing the diffraction of an ultra-short-wave radiation around
the earth and the amplitude of the direct-diffracted component can
be calculated at once.
That is, it can be calculated at once if the air mass has no refractive
bending effect upon the radiation trajectory. Since such a bending
effect is certainly present at times, and is equally certainly variable,
even if only slowly, it must be taken into account.
508
BELL SYSTEM TECHNICAL JOURNAL
If the refractive index of the air varies as a power of the distance to
the earth's center, it has been shown ^ that the actual state of affairs
can be duplicated by a homogeneous atmosphere over an earth, the
radius of curvature of which is greater than that of the actual earth
and is calculable from the exponent of the height variation function.
With this "effective" earth radius, the formulae already mentioned
become usable. If the air refractive index does not vary as a power of
the distance to the center of the earth we must take that exponent
which gives the best first order fit over the height covering the re-
fracted wave front, the alternative being a prohibitive complication of
the theory.
A plausible physical picture of the fading mechanism can now be
set up. If we lump the four boundary reflected components in one,
and plot as a function of the distance, we have curves "yl " of Fig. 15.
REFLECTING BOUNDARY
Fig. 14 — Drawing illustrating the four components of a single reflection
at an air boundary.
Curves "5" are the Wwedensky ^ * and Gray ^ theories. These are
for our Highlands-East Moriches circuit with the average effective
earth radius of 8500 kilometers and a 1500-meter boundary height.
If we now imagine a receiver moving away from the transmitter we
shall first traverse the zone of high "5" amplitude with no fading
present. The signal amplitude will, for any given near-by point, and
for any given antenna ampere-meters, depend on the height of the
antenna above the ground and the ground constants. As the distance
to the transmitter increases, the falling "5" curve approaches the
rising "^" curve in ordinate and we enter a disturbed region where,
for any instability of the boundary, more or less complete interference
can result and fading will occur. (One such instability occurs when
* There is an error in the formula, as given by Wwedensky. It is corrected here.
See appendix II.
ULTRA-SHORT-WAVE TRANSAilSSION
509
a boundary with an irregular surface is carried past the reflection zone
by the normal motion of the atmosphere.) A further increase in
transmitter distance and the "5" or residual curve drops out of the
picture leaving only the "A'' curve and, presumably, fairly steady
signal amplitude conditions. The location of these zones of undis-
turbed and disturbed signal will vary from day to day as: (1) the
reflection coeflacient and height of the layer change, (2) the effective
radius of the earth changes. The effect of the height of the layer is
shown in Fig. 16.
OQ 20
10
-10
-20
\
k.
\
\
^
"Bv REFRACTED -
^ DIFFRACTED
>
<;b'h
X
s.
\,
s
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1
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REFLECTED Ns
s
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1
^
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S
N,
- 1
f
1
r
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s...,.
60 80 100 120 140 160
DISTANCE FROM TRANSMITTER IN KILOMETERS
180
Fig. 15 — Calculated curves for air boundary reflected and earth refracted-
diffracted radiation components, in both vertical and horizontal polarization.
Short doublet antennas, 1 kw. power radiated, wave-length 4.7 meters, o- = 5 X 10~^^
(E.M.U.) and e = 80 for sea water. Height of transmitter, antenna 42 meters, of
receiver antenna 5 meters, air boundary height 1500 meters, effective radius of earth
8500 kilometers.
Since the major lobe of the polar characteristic of any simple an-
tenna, such as ours, is directed forward and away from the earth, the
signal intensity at the reflecting boundary surface will be comparatively
high and will, in some measure, make up for a small reflection coeffi-
cient. For longer waves, such as broadcast waves, the high level of the
" 5 " curve will move the disturbed zone so far out that the low residual
signal level and the Kennelley-Heaviside layer reflections will conceal
510
BELL SYSTEM TECHNICAL JOURNAL
or mask the atmospheric boundary reflections. Several observations
which can be ascribed to such boundaries have nevertheless been
published.''- ^ Obviously, only boundaries lying considerably higher
than those discussed here will give the path differences to produce the
same type fading at these longer waves. At the same time the ap-
parent diffuseness of a boundary will fall off with increase in wave-
length, thus removing the restriction of reflection to near grazing
incidence angles only.
60
■^
SH
/
/
c
A6= C10)"5
Y-
s^e = (io)-'*
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.18 3.2 5.6,
Ae=(Xior^
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80 70 60 50 40 30 20
RESULTANT FIELD, DECIBELS BELOW FREE SPACE
Fig. 16 — Calculated field strength curves showing the effect of air boundary height
and density on the reflected radiation component, for the Highlands East Moriches
circuit. Transmission path 112 kilometers, over sea water, wave-length 4.7 meters,
polarization both vertical and horizontal. Vertical antennas 42 and 5 meters high,
horizontal antennas 45 and 9.5 meters high, respectively.
This tentative mechanism also explains several other observed
features. Thus, for a given type of boundary instability, the fading
rate will increase as the wave-length decreases. Furthermore, since
the slope of the "5" curve increases as the wave-length decreases, the
ULTRA-SHORT-WAVE TRANSMISSION 511
disturbance zone is effectively moved nearer the transmitter and the
probability of increase in fading amplitude is enhanced. The usual
increase of fading with decrease in wave-length is thus explained.
When the wave-length difference is small, on the other hand, the fading
type should be much the same on both wave-lengths, as was generally
found. The lack of coincidence would arise from the fact that the path
difference being a considerable number of wave-lengths, a small wave-
length change can introduce a marked randomness in fading without
appreciably affecting the type.
As has been mentioned earlier, a multiple of reflecting boundaries is
the normal condition, rather than that of a single boundary. This
circumstance, without invalidating the explanations already given,
makes a further elaboration of the theory possible. The "roller"
type or component of fading, in particular, requires explanation. In
addition to the smooth signal modulation, from which the name has
been derived, this type of fading is characterized by showing more or
less frequent deep minima or drop-outs and these are often distinctively
twinned. Further, the roller component is that component of fading
which shows coincidence, in spite of wave-length or polarization dif-
ferences. Such coincidence indicates small path difference and this
is what we have when a double boundary or stratum exists. Such a
stratum would give two "A" components and, if of variable thickness,
would, as it was carried along by the prevailing air currents, give the
steep, deep, minima at phase opposition thickness. Further, if the
stratum contour were that of a hump, thick enough to carry the second
"A" component past phase opposition to the first one, the twinned
minima would result as the hump entered and left the reflection zone.
Occasionally the two "A " components would add properly, with the
residual "B" component, to give complete extinction, a result less likely
from the phase addition of a single "^" and the "5" component.
This explanation of "roller" fading assumes, tacitly, that the "B"
component is, at the time, relatively subdued, that is, the disturbance
zone has moved inwards due to an increase in the reflection coefficient
of the layers or to a decreased "effective" earth radius. The fine
structure often appearing at the bottom of a prolonged roller minimum
corroborates this, the mutual cancellation of the two "A" components
having uncovered, so to speak, the weaker "B" component with its
much shorter traversed path.
With the "roller" condition characteristic of high "A" component
signal amplitude, the "scintillation" condition would be characteristic
of low "A" component signal amplitude, the relatively steady "B"
512 BELL SYSTEM TECHNICAL JOURNAL
component having superposed on it a small amplitude, variable phase,
"yl" component. A relatively low mean amplitude value and the
coincidence of scintillation conditions with conditions of convective
instability of the atmosphere would thus be explained. All the
scintillation records came on days of relatively high wind and convec-
tive instability. A turbulent atmospheric condition would dissipate
or attenuate any boundaries, especially the lower ones. The rapid
flutter about the mean amplitude value is the normal expectation from
a high, turbulent, low reflection coefficient boundary.
Our two polarization results are qualitatively explicable on the
mechanism proposed. As can be seen in Fig. 15, the change from
vertical to horizontal polarization results in a relative lowering of the
"5" curve without much change in the "^" curve, which should
result in increased fading. For our circuit and a boundary at 1500
meters the relative "5" vs. "yl " drop is 13 db.
As Fig. 16 shows, the variations of the "^ " components with height
are markedly different for the two polarizations. The "Ay" compo-
nent falls steadily with height up to 4700 meters; the "Ah'' component
has a deep and sharp minimum at 3000 meters after which it rises
again. Since most of our observations concerned boundaries at 2000
meters or less, this high altitude disparity between "Ah" and "Ay"
does not affect our explanation. The disparity between vertical and
horizontal fading should be much more marked for high boundaries
than for low boundaries.
Further Experimental Curves and Checks
The curves given have illustrated the variability in the fading, a
variability which no short period of recording can encompass. The
tentative explanations proposed have been shown to be in accord with
several of the features characteristic of this fading. Certain other
experimental results will now be adduced which offer further verifica-
tion along somewhat different lines.
For the forty-five mornings on which simultaneous recording was
carried out during the United States Weather Bureau plane flight, we
have calculated, from the airplane data, the values of the "A" and
"B" components. As stated earlier, there were twenty-four days when
boundaries above 400 meters altitude, and of sufficient distinctness to
be fairly accurately estimated (Ae = 10~^) were shown by the meteoro-
graph records. For these the "A" components have been computed.
By taking the dielectric constant gradient for the first half kilometer,
the effective earth's radius was determined and inserted in the Wwe-
densky formula to give the "5 " component. These calculated values
I
ULTRA-SHORT-WAVE TRANSMISSION
513
("yl " component as triangles, "5" component as circles) are plotted
on Fig. 17 together with the maximum and median * observed values.
These latter are joined by lines. For the 10 mornings on which no
boundaries were evident the calculated "5" component appears to
be some 8 db higher than the observed values. With this correction
the agreement between observed and total calculated fields is fairly
good. A partial explanation of this 8 db discrepancy may lie in the
fact that the ocean water trajectory assumed in the calculation differs
from the actual one by the land terminals and the three tongues of land
intervening.
1 - BOUNDARY EVIDENT
AC* tO-S
2 -NO BOUNDARY
3 - SMALL BOUNDARY
AetlO"^
4 - INDEFINITE BOUNDARY
6- POSSIBLE BOUNDARY
BELOW 400 METERS
-M*-3-
a 50
-c^i
o °o no4>°0 $
tr^
a — O-Q-
_J I I I I I L
- rf) -^ to fO
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4- CALC. "A" COMPONENT r^-OBSERVED MAXIMUM
0-CALC."B" COMPONENT L— OBSERVED MEDIAN
•LAKEHURST DATA "A"
Fig. 17 — Comparison of "^" and "S" radiation components, calculated from
the U. S. Weather Bureau free air data, with measured maximum and median signal
strengths. Vertically polarized transmission.
On the twenty-six morning frequency sweep runs there were only
three on which the predominant sweep pattern was simple enough to be
interpreted as due to two components with path difference greater
than 75 meters. For those days a series of measurements of the film
patterns was made by determining the frequency spacing between
a maximum and a minimum and calculating the resulting path dif-
ference and boundary height. The dielectric constant-height function
was also calculated from the Weather Bureau data. These curves are
* The signal is half of the time greater and half of the time less than its median
value. For random phase with "S" component equal to "A" component the
resultant median value signal is V2 X ^ or 3 db up; it falls from this value to "A "
as "B" decreases to zero.
514
BELL SYSTEM TECHNICAL JOURNAL
plotted in Fig. 18 with the calculated boundary heights set down at
the right hand, spread out in time of observation. The boundary
height coincidence is pretty definitely located in this manner. Many
of the more complicated frequency sweep patterns carried a fine struc-
JUNE 23, 1936
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TEMP IN ° C DIELECTRIC CONSTANT
2 6 10 14 18
VAPOR PRESSURE
MILLIBARS P^
600 6 30 700
AM EASTERN STANDARD TIME
Fig. 18 — Comparison of boundary heights shown by the U. S. Weather Bureau
free air data, with boundary heights measured from frequency sweep patterns.
Horizontally polarized transmission.
ULTRA-SHORT-WAVE TRANSMISSION 515
ture which indicated weak boundaries at higher altitudes up to, roughly,
5.5 km.; most of the patterns, however, were characteristic of layers
below two kilometers. The path difference corresponding to two kilo-
meters is 85 meters. The theoretical limit of resolution of the amplifier
band for a maximum to minimum frequency spacing is Al — 2(C/A/)
where Al = path difference, C = velocity of light and A/ = frequency
band. For A/ = 2 X 10^ cycles this gives 75 meters, and hence
boundary heights at and below 1900 meters are unresolvable by our
receiver. It is a remarkable result that the bulk of the disturbing
boundaries should lie so low.
It was mentioned earlier that several observations referable to air-
mass boundaries have been published. In addition there have been
reports, for three consecutive years, of long distance ultra-short-wave
reception by American amateurs ^ during the month of May. We
have copies of the U. S. Weather Bureau atmosphere cross-sections
for several of these days and have been curious enough to examine
them. On May 9, 1936, during the long distance amateur reception,
there was an extensive boundary at 4 km. between an upper Superior
air mass and a lower Tropical Gulf air mass. On May 15, 1937, a
similar boundary at 4-5 km. had a Superior air mass above a wedge of
Transitional Polar to Tropical Atlantic air. Below this at 3-4 km. lay
a Transitional Polar Continental air mass.
On June 11, 1936, when Colwell and Friend^ report an extra strong
0-2 km. " C" reflection, a subsiding Transitional Polar Pacific air mass
lay above a Transitional Polar Continental air mass with the boundary
at about 1.5 km. On June 29, 1936, when they reported a very strong
3.5 km. "C" reflection, there existed four wedge-shaped air masses
with a Superior air mass over a Transitional Polar air mass at 3-4
kilometers. The wave-lengths used were 186, 125 and 86 meters
approximately.
These coincidences may or may not be significant but it is very
questionable that any boundaries at such altitudes are due to either
electron or gas ion distributions.
The characteristic properties of North American air masses have
been published," as average summer and winter values, and show some
marked seasonal differences. The greater dielectric constants for
summer conditions are due chiefly to greater water content.
For a single air-mass distribution, horizontal stratifications are at a
minimum and the radio transmission is via the "B" component.
This component can be calculated from the corresponding effective
earth radius. The table below gives this radius for three important
air mass types.
516
BELL SYSTEM TECHNICAL JOURNAL
Effective Earth. Radius
Air Mass Type
Summer
Winter
Tropical Gulf- — Tg
Polar Continental — Pc- ■ .
Superior — 5
. . 1.53 X R
.. 1.31 X R
.. 1.25 X R
1.43 X R
1.25 X R
1.25 X R
" R" = actual earth radius
The boundaries between different air-mass types furnish discon-
tinuities adequate for radio reflections. The greater the stabiUty
of the boundary, the more abrupt it is Ukely to be. In general, when
"5" air overlays either "T^" or ''Pc'' air, the resulting boundary is
stable. Possible discontinuities, for the three types discussed, may be
summarized in the following table. Here the positive sign means that
the radiation originates in the more refractive medium. For stability
the lower medium is the denser though not necessarily the more
refractive.
Ae X 106
Altitude
Summer
Winter
SITg
SIPc
TglPc
SITg
SIPc
TglPc
1.0 Km
100
50
30
20
10
10
-80
-40
-20
55
50
35
25
15
10
-30
2.0 Km
3.0 Km
-35
-25
Concluding Remarks ^
The characteristics of this seventy -mile circuit indicate that for
ultra-short-wave transmission it rates as a long distance one. If we
assume that the air refraction is on the average such that the effective
earth's radius is 4/3 the actual one, then the receiving station lay 1400
feet below the line of sight from the transmitter. This is equivalent to
0.57° below the horizon. The reception, using high efifective-height
antennas, was good; there was, however, very little lee-way left, above
set noise, for reception with simple doublet antennas. Any longer
circuit will require to be terminated on elevations such as to keep the
intermediate horizon height down. The fading was too slow to be
noticeable on amplitude modulated speech unless a deep minimum or
drop-out occurred.
The circuit was probably unusable for television, most of the time.
A system adhering to the R.M.A. standard ^- of 441 lines on an inter-
ULTRA-SHORT-WAVE TRANSMISSION 517
laced 60-cycle scanning will have a unit time element of 0.17 micro-
seconds. This corresponds to a path difference of 51 meters and only
a fraction of this is necessary to produce a ghost. A rough estimate of
the boundary height range involved in our fading is one-half to five-and-
one-half kilometers. The corresponding path difference range is 8 to
580 meters. As the fading records show, no matter whether the ''A"
or the "B" component predominated, the other component was
usually present in amplitude only second to the other. It may be
pointed out that where a standing wave system exists, ^"^ reflected
components with much larger path differences than those recorded
here are almost certain to be found.
Appendix I
In the Wwedensky * paper the author applies his theory to one of the
experimental curves from a previous paper of ours. He uses the nor-
mal earth radius " R," however, without any correction for air re-
fraction. If we assume, as a more probable effective earth radius, the
value 4/3R,^ the agreement with our curve is markedly improved.
Appendix II
In the first Wwedensky paper, Tech. Phys. U. S. S. R. Vol. 2, p. 632,
1935 eq. (7, 1) the sign of the term Ir^p sin 2dm should be minus.
Appendix III
The fading produced by moving bodies such as airplanes has been
referred to in one of our earlier papers.^" It happened one day, during
the present investigation, that fading of this type appeared when
mechanical recorders were being used and, by speeding up the paper, a
record in two polarizations was obtained. The airplane itself (or other
cause) was not visible. The results are given in Fig. 19. Again the
horizontal component was the worse one. At first the two fadings,
both fine and coarse components, were in step; later they passed en-
tirely out of step where the fading was so rapid as to smear the paper.
These "airplane" fadings were observed, off and on, at other times
but were not recorded.
References
1. Englund, Crawford and Mumford, Bell System Technical Journal, Vol. 14, p. 369,
1935.
2. Brown and Leitch, Proc. I. R. E., Vol. 25, p. 583, 1937; Norton, Proc. I. R. E.,
Vol. 26, p. 115, 1938.
3. Ross Hull, Q.S.T., Vol. 21, p. 16, 1937, May.
4. B. Wwedensky, Tech. Phys. U. S. S. R., Vol. 2, p. 624, 1935; Vol. 3, p. 915, 1936;
Vol. 4, p. 579, 1937.
518
BELL SYSTEM TECHNICAL JOURNAL
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ULTRA-SHORT-WAVE TRANSMISSION 519
B. van der Pol and Bremmer, Phil. Mag., Vol. 24, p. 141, 1937; Vol. 24, p. 825,
1937.
5. Miss M. C. Gray, paper to be published.*
6. Schelleng, Burrows and Ferrell, Froc. I. R. E., Vol. 21, p. 427, 1933.
7. Colwell and Friend, Nature, Vol. 137, p. 782, 1936; Phys. Rev., Vol. 50, p. 632,
1936; Colwell, Friend, Hall and Hill, Nature, Vol. 138, p. 245, 1936; Friend
and Colwell, Proc. I. R. E., Vol. 25, p. 1531, 1937.
8. Watson Watt, Wilkins and Bowen, Proc. Roy. Soc, A, Vol. 161, p. 181, 1937.
9. Q.S.T., Vol. 21, p. 27, 1937, July.
10. Englund, Crawford and Mumford, Proc. I. R. E., Vol. 21, p. 464, 1933.
11. H. C. Willett, Bull. Amer. Meteor. Soc, Vol. 17, p. 213, 1936.
12. Beal, Television, Vol. 2, p. 15, 1937, R.C.A. Inst's. Press.
13. Englund, Crawford and Mumford, Nature, Vol. 137, p. 743, 1936.
* The case of vertical polarization is treated by references 4, that of horizontal
polarization by reference 5.
Amplitude Range Control
By S. B. WRIGHT
The art of controlling the amplitude range of telephone signals
involves recognition of certain characteristics in addition to those
used to specify the performance of ordinary transducers. Funda-
mentally, three kinds of characteristics are necessary to distinguish
different range control devices. They are (1) the steady-state
input-output characteristics, (2) the time actions, and (3) the range
over which they function. In some cases, several secondary char-
acteristics may be of interest, but they need not be considered in
determining to which class a particular device belongs.
This paper discusses and classifies these characteristics.
Introduction
TN a "non-linear" transducer, the output power is not proportional
-■- to the input power. Consequently, the ratio of maximum to
minimum power at the output differs from that at the input. But the
ratio of maximum to minimum power is an expression of amplitude
range. A device designed to alter this ratio may be called a range
controller.
In telephony the term range controller includes many devices ^ having
specific names, such as limiters, volume control devices, range reducers,
compressors, vogads, expandors, etc. These devices have many prop-
erties in common with telephone repeaters, and a repeater may be
considered as a special case in which any non-linearity which may
exist between the output and input is unintentional.
The purpose of one type of range controller is to reduce the range of
significant intensities of signals applied to a telephone circuit so as to
ease the requirements of the transmission medium with respect to
overloading and noise interference. Such a device is placed at the
transmitting end of the circuit. When the range is compressed at the
sending end of the circuit it may sometimes be desirable to expand it
at the receiving end to the original range. This is done with a device
having, in general, the same dynamic characteristic as the compressing
device, but a range change which is complementary. The purpose of
the expandor is to reduce the noise heard by the listener as well as to
compensate for whatever characteristic signal modification occurred in
^ For numbered references, see end of text.
520
AMPLITUDE RANGE CONTROL 521
the process of compressing the original wave. Sometimes an expandor
is used at the receiving end to reduce the gain in the intervals between
the main signals even when no compressor is employed. This is an
example of using a range controller to correct defects in the medium.
As is well known, the performance of a repeater is specified by such
characteristics as impedance, amplification, frequency band, noise,
and output carrying capacity. The performance of a non-linear device
involves some additional characteristics. The primary ones are (1)
the slope of the input-output curve, which tells how the range is
changed, (2) the dynamic operation, which tells the manner in which
the output varies with time following a given change in input, and (3)
the range, which tells the region over which the device exercises control.
It may be helpful to imagine a range controller as an amplifier in
tandem with an adjustable attenuator, the loss of which may be
changed either instantly or slowly to follow in some predetermined
fashion changes in the signal. For simultaneous operation, this device
could put out a wave which is a simple function of the input, but if
the operation were delayed by a definite interval the device would be
required to respond in a complex fashion in accordance with a re-
collection of what had occurred in the signal during the delay period.
Such delayed adjustment would be very crude for intervals comparable
with the periods of fundamental speech frequencies. To obtain
practical regulation of the delayed type it is necessary to increase the
delay beyond this range and base the control upon the amplitudes of
the syllables. When the delay is increased to a point where it is
comparable with the syllabic periods its usefulness is again reduced.
Part 1 — Control Ratio
Fundamental Characteristics
Figure 1 shows how waves may be altered by a device having a
given output-input characteristic, assuming the operation is instan-
taneous. As this figure is plotted on a db scale, only the stronger
portions of positive values of the wave are shown. A similar diagram
could be drawn for negative values. The output-input characteristic,
although a straight line in this kind of diagram, would of course be
parabola-like if plotted on a current or voltage basis. By selecting
points, such as A (or B), on the input wave and determining the rela-
tive outputs A" (or B"), the corresponding resultant wave is obtained.
In this case, the resultant has a flatter top than the original sine wave,
and this illustrates the capabilities of the device in increasing weak
signals with respect to the strong ones and also suggests that distortion
522
BELL SYSTEM TECHNICAL JOURNAL
may accompany the transformation. Such effects depend upon the
slope of the output-input characteristic.
The control ratio of a range controller might be defined as the output
range in db divided by the input range in db within the non-linear
region of interest. The ratio is obtained in such a way as to eliminate
transient effects, i.e., using steady-state sine waves.
Typical Control Ratios
Figure 2 shows some typical output-input characteristics for various
transducers having control ratios between zero and infinity. While
INPUT IN DECIBELS BELOW ARBITRARY REFERENCE
60 50 40 30 20 10 0,
COMPRESSED
(OUTPUT) WAVE
Fig. 1 — ^The signal modification caused by a non-linear transducer depends upon
the slope of the output-input characteristic.
these typical characteristics are straight lines there is nothing to pre-
vent a range controller having a control ratio which varies with input.
However, when complementary action is required at the receiving end
it is more readily obtained when the control ratio is constant. Also,
some physical elements used in the design of range controllers are
most readily adapted to a straight line characteristic.
Compressors (that is, devices having control ratios less than 1) may
be divided into two classes: (1) Complete * and (2) Incomplete. In a
complete compressor (control ratio = 0) the output is held constant
within the range of the device. This control ratio gives a maximum
* This is not usually of practical interest but is useful as an ideal limit of operation.
AMPLITUDE RANGE CONTROL
523
of possible noise improvement and also a maximum of signal modifica-
tion. There is, however, no information in the compressed signal
INVERSE
COMPLETE
EXPANDOR
IDEAL TRANSDUCER,
COMPLETE REPEATER OR
COMPRESSOR ATTENUATOR
COMPLETE
EXPANDOR
-RANGE INVERTERS -
♦-COMPRESSORS-
-EXPANDORS-
-45 0 45
ANGLE OF SLOPE IN DEGREES (FROM HORIZONTAL)
(b)
Fig. 2 — If transducers are classified with respect to the slope of the input-output
characteristics, several fields of action with definite demarcations result.
which would serve to indicate how much compression occurred.
Consequently, if it were desired to restore the original range, it would
be necessary to transmit this information in addition to the compressed
524 BELL SYSTEM TECHNICAL JOURNAL
signal. The gain of the restoring device would be guided by this
auxiUary information. Hence, the device used to pass the information
along is called a "pilot channel." Various types of pilot channels are
listed in Part 4 as secondary characteristics of the control.
When the control ratio is between 0 and 1 the compression is in-
complete. A wave compressed in this manner has the property of
being able to cause re-expansion at the receiving end since the output
amplitude bears a definite relation to the original, assuming constant
transmission over the intermediate circuit.
In the field of expandors having a control ratio between one and
infinity the signal modification is opposite to that of compressors.
Thus a convenient method is available for restoring the original wave
shape by using an expandor having a control ratio which is the re-
ciprocal of that of the compressor at the sending end.
Effects of Control Ratio
The control ratio is useful in determining the effectiveness of a
device in improving transmission in the presence of noise in the med-
ium. When noise alone is acting on the device, the noise determines
the action in a manner similar to speech. When both noise and speech
are present, the action is determined by the sum of the two. Thus,
room noise applied with the speech will be compressed or expanded
exactly as if it were part of the speech. In the case of a compressor
used at the sending end of a noisy circuit, an input range of say 60 db
might be compressed to 20 db, by using a control ratio of 1/3 over the
entire input range. At a point where the strongest signals are un-
changed, the weaker signals would then be 40 db stronger than when
the compressor was omitted. The improvement of the signal and
applied noise with respect to noise in the medium thus depends on the
difference in ranges at the input and output which depends on the
control ratio.
A large part of the usefulness of an expandor is in changing the
apparent ratio of speech to the noise heard in the absence of speech,
since the noise is generally weaker than speech and is made even less
compared to speech by expansion. This is in spite of the fact that at
any instant the signal-to-noise ratio is the same at the output as at the
input. When the noise is comparable with the speech in amplitude,
or when the noise is so weak as to be negligible without a controller,
there can be no improvement in the noise conditions in using these
devices. Between these two limits, the noise improvement rises to a
maximum value also determined by the control ratio, and the time
actions and range to be discussed.
I
AMPLITUDE RANGE CONTROL 525
A receiving range controller also changes variations in the trans-
mission medium in proportion to the control ratio.
Part 2 — Time Actions
Instantaneous Control
A device having a given control ratio might have its gain changed
simultaneously with the applied e.m.f. The signal modification
would become greater as the control ratio departed farther from unity
and the modified signals would approach rectangular wave shapes
at the limiting control ratios. Unless instantaneous compression is
limited to a very small part of the signal range, an incomplete in-
stantaneous expandor (inverse rooter) is required at the distant end
which does the reverse of what is done at the transmitting end to
restore the signal to substantially its original form. Due to the
characteristics of the compressed signals, however, a transmission
bandwidth without appreciable amplitude or phase distortion of two
to three times the normal is necessary for high quality transmission.
Rectified Control
To avoid the necessity of transmitting such a wide band of fre-
quencies, as well as to permit the use of a single device without restor-
ing, in which case the distortion is limited to a value which is permis-
sible from the standpoint of a listener, practical devices do not operate
instantaneously. Instead, the gain is controlled by the charge on a
condenser, which is controlled by rectified waves. The action of such
an arrangement will now be discussed.
Consider a wave formed by subtracting two sine waves equal in
amplitude, one having a frequency 10 per cent less than the other.*
A portion of such a wave is shown in Fig. 3a. This wave is equivalent
to a cosine wave of frequency one-half the sum of the two frequencies,
as shown by the instantaneous voltages of Fig. ?>a, multiplied by a
secondary wave (envelope) of frequency one-half the difference of the
two original frequencies.
The instantaneous voltages of the wave of Fig. 3a vary from a
positive maximum through zero to a negative maximum. Curve a of
Fig. 4 is a summation of most of the instantaneous e.m.f. 's of Fig. 3a
with respect to their occurrence. About 99 per cent of the instantane-
ous voltages are in the ranges shown, the remainder being in the range
between the upper and lower halves of Fig. 4.
* This illustration is not directly comparable with speech, but it contains some of
the' attributes which are comparable in this analysis, besides being readily repro-
ducible and relatively simple.
526
BELL SYSTEM TECHNICAL JOURNAL
Figure 3b indicates values for the same wave in which the negative
ordinates have their signs reversed by means of an ideal full-wave
rectifier. The resulting wave contains frequencies which were not
present in the original, prominent among them being second and higher
harmonics of the original. The range of instantaneous values shown
5 0
RECTIFIED ENVELOPE-^
RECTIFIED
TIME
Fig. 3 — A wave's amplitude varies from positive maximum to negative maximum.
If symmetrical, the amplitude may be expressed as varying in only one direction from
zero to maximum by rectification.
on Curve b of Fig. 4 is only half that of the instantaneous voltages.
About 99 per cent of the values lie in a 60 db range.
The instantaneous values of the envelope of the rectified wave
follow curves 3c and 4c. In speech the envelope is composed of
many rather low frequencies which are determined by the rates of
enunciation of syllables. For this reason they are sometimes called
the syllabic frequencies. If it were possible to make the control vary
as a function of the envelope, the result of using a control ratio of 1/2
on the wave of Fig. 3a would be as shown in Fig. 5c. This was
I
AMPLITUDE RANGE CONTROL
527
obtained by multiplying the original wave by a factor which is inversely
proportional to the rectified envelope. For comparison, the original
wave is shown in Fig. 5a, and the result of instantaneous compression
I
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10 20 30 40 50 60 70 80 90 100
PER CENT OF TIME EMF IS ETQUAL TO OR LESS THAN EMF SHOWN
-The amplitude range of the wave of Fig. 3 is infinite on a db scale but most
of the values are bunched in a much smaller range.
by the same control ratio in Fig. 56. It is assumed that the arbitrary
reference voltage which is not changed by compression corresponds to
the maximum value of the input wave, although any other value
might be used instead. It is evident from Fig. 5 that envelope com-
528
BELL SYSTEM TECHNICAL JOURNAL
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AMPLITUDE RANGE CONTROL 529
pression would result in less distortion than instantaneous compression.
The extra frequencies formed that were not present in the original
wave are the envelope frequencies, so that the additional band required
to transmit this wave faithfully is negligible.
Dynamic Operation
The measurements ^ and adjustments of speech amplitudes in com-
mon use are made with devices that integrate the effects of the wave
over certain time intervals. They do this in a rather complicated
manner, however, so that it is difihcult to express the resulting quanti-
ties in terms that are generally understood.
In the measuring instruments the rectified voltages are impressed
on a condenser before being sent through a meter. The readings of
the meter are, therefore, proportional to the voltage on the condenser
modified by the damping of the meter. The voltage is made up of the
sum of the elTects of all the instantaneous voltages that have been
applied to the condenser from the beginning of time to the instant under
consideration. These effects die out so rapidly, however, that the
instantaneous voltage on the condenser is practically determined by the
voltages received in the immediate past. The condenser may be said
to have a memory but a short one. In range control devices, the
condenser forms the voltage which determines the amplification of the
device.
To distinguish this voltage on the condenser from the applied voltage
at any instant, we may call the former an "impression" of the original
wave. If the time constant RC is small we get strong impressions
similar to the rectified applied wave and its envelope, and if it is large
we get weak impressions quite different from the applied wave but
something like the rectified envelope.
Figure 6 shows the impressions of the wave of Fig. 3a, using four
different values of time constant RC as compared to P, the period *
of the envelope. Figure 7 shows smoothed summation curves of the
impressions of Fig. 6 formed during the time P/2. Comparing this
with Fig. 4, it is evident that the "bunching" effect for the distribution
of impressions is largely between those for the rectified instantaneous
and envelope curves. For the longer time constants, i.e., weak im-
pressions, this is not the case for the weaker e.m.f.'s.
* This is twice the duration of Fig. 3, since only half a cycle is illustrated. It is
assumed that C is completely discharged at the time this wave is applied. In prac-
tice, the rectifier impedance varies with the applied e.m.f. so that the results are
not as simple as in this illustration. In general, the time actions are different de-
pending on whether the applied e.m.f. is increasing or decreasing.
530
BELL SYSTEM TECHNICAL JOURNAL
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AMPLITUDE RANGE CONTROL
531
Referring again to Fig. 6, it will be seen that for the two smaller
values of RCjP the impression curves are composed of (1) the envelope
frequency, (2) double the fundamental frequency, and (3) a small delay
which can often be neglected. An approximation to envelope com-
pression is therefore possible by choosing RCjP to be in the proper
range, i.e., .0025 to .025, and making the output vary as a root or power
ot the impressions thus formed.
Figure 8 shows the result of compressing the wave of Fig. Za by
using the impressions of Fig. 6 to determine the amplification. It was
0 10 20 30 40 50 60 70 80 90 100
PER CENT OF TIME IMPRESSION IS EQUAL TO OR LESS THAN IMPRESSION SHOWN
Fig. 7 — The amplitude ranges of the impressions shown in Fig. 6 are bunched
differently, depending on the time constant. A "volume" measurement means that
a given impression is exceeded a small percentage of the time. In speech the peaks
are relatively higher than in the wave illustrated.
assumed that the amplification varies in inverse proportion to the
square root of the impression. The resulting waves for RCjP = .0025
and .025 (medium impressions) are recognizable as something like the
original wave. However, for the larger values of RCjP (weak im-
pressions), the distortion at the beginning of the wave is quite large.
This is because the impressions are formed so slowly that a longer time
is required to drive the gain down to the desired value.
In order to compare impression compression with instantaneous
compression, the ordinates of Fig. 5 and 8 were plotted in Fig. 9.
This shows that the greatest possibilities of bunching the waves into
a narrow range result from the use of instantaneous compression {b),
since the ratio between any value and the maximum is modified by the
532
BELL SYSTEM TECHNICAL JOURNAL
c o
o o
AMPLITUDE RANGE CONTROL
533
a
INPUT,
UNCOMPRESSED
OUTPUT, COMPRESSION
BY IMPRESSIONS
• OUTPUT, ENVELjOPE
COMPRESSION
>0 60 70 60 90
PER CENT OF TIME EMF IS EQUAL TO OR LESS THAN EMF SHOWN
(POSITIVE VALUES)
Fig. 9 — The amplitude ranges of the compressed waves of Fig. 8 are shown,
together with those of Fig. 5. The amount of signal modification (and noise im-
provement) for any amplitude below maximum may be correlated readily with the
time constant.
534 BELL SYSTEM TECHNICAL JOURNAL
control ratio. In the case of envelope compression (c), the lag causes
a reduction in the amount of compression of the instantaneous voltages
and the result is seen to be about half way between curves a and h of
Fig. 9. The remaining curves of Fig. 9 are representative of the
device ^ used on the long-wave transatlantic radiotelephone circuit.
Volume Control
To avoid both a large range and also the necessit}^ for a continuous
record, practical speech amplitude measuring instruments are not
directly concerned with either instantaneous, envelope, or impression
voltages. Instead a value is determined, corresponding to an im-
pression which is exceeded only a small percentage of the time. This
is the principle underlying speech measurements with "volume in-
dicator" type of instruments. In the case of speech, which is much
more complex than the simple wave we have discussed, curves like
Fig. 9 are steeper, i.e., there are relatively more peaks and a larger range
to complicate the problem.
A particular device capable of compressing according to the re-
quirement that the dynamic "volume" range should be reduced, is
attained by a combination of several separate range controllers. One
is provided to reduce the gain very rapidly when the output volume is
too high. A second increases the gain, at a much slower rate, when the
impressions formed on the condenser are consistently too low. A
third disconnects the condenser from the input when the applied
voltage is very small, so that the distortions inherent in change of gain
by weak impressions will occur only at times of large and sudden
decreases of volume. In the device ^ employed to control volumes
applied to a radio transmitter at Norfolk, Virginia, a fourth control
provides for rapid partial compression of high peaks, thus improving
the modulation. It is unnecessary to re-expand for the purpose of
restoring the intelligibility, since the distortion is virtually limited to a
change in loudness.
Part 3 — Range
Range controllers, like repeaters and attenuators, are limited as to
the input range they can accept and the output range they can provide.
These limits may be due to thermal noise at the low end and output
carrying capacity at the high end. Heretofore, in this paper, the
terms "input" and "output" have been purposely left somewhat
vague so as to be as general as possible. However, the limits of input
and output of a range controller take on particular significance when
it is considered that the signal input range may difTer both from the
AMPLITUDE RANGE CONTROL 535
input range of the device, and also from the range over which control
is exercised.
This control range may be defined as the difference between the
maximum and minimum values of an applied wave over which a
device is designed to function in a specific non-linear manner. It is
usually expressed in db, and may apply to any measure of the applied
signal, such as instantaneous voltage, rms steady-state sine waves, or
a dynamic measure such as volume. The values dividing the con-
trolled range from the uncontrolled ranges may be referred to as the
"control points."
Certain advantages in some cases have been found from restricting
the control range. This is accomplished by placing one or both of the
control points inside the useful amplitude range. The position of the
control point may be moved arbitrarily over a wide range by putting
an ordinary repeater (or attenuator) in tandem with the range con-
troller. A given amount of compression at the high amplitude end of
the range gives a real signal-to-noise advantage for a much greater
proportion of applied e.m.f.'s than the same amount of compression
at the low end of the range. In either case the distortion would be less
than that of a full range compressor. When expandors with limited
range are used, they are subject to the limitation that variations in the
medium are increased, but to a lesser extent than full range expandors.
Part 4 — Classification of Range Controllers —
Secondary Characteristics
Table I, page 536, suggests how the conceptions of control ratio,
time actions and range already discussed might be employed to dis-
tinguish a variety of devices. In cases where more than one device
is covered by a given control ratio and time action the distinction is
that the ranges are different. The names of devices used in this table
are those which have been used in the past to distinguish the devices
one from another.
Nomenclature
Using the above conceptions of the three primary characteristics,
it has been found possible to devise a notation to distinguish all the
known devices in this field. As an example of how this proposed
system of nomenclature would be applied. Table II, page 537, gives
three columns. Column 1 sets forth the arbitrary names that have
been used in the past to distinguish certain devices which have come
into use. Column 2 gives descriptive names which specify the three
fundamental characteristics. In column 3 each device is named by
three symbols defining the three fundamental characteristics, and a
536
BELL SYSTEM TECHNICAL JOURNAL
TABLE I
Classification of Range Controllers
Typical
Compressors (r < 1)
Expandors (r > 1)
1 ime Actions
Full Range
Limited Range
Full Range
Limited Range
Instantaneous
Rooter
Peak Chopper,
Voltage
Limiter
Inverse Rooter,
(Squarer)
Voice Operated
Relays, Cross-
talk Sup-
pressor
Syllabic
Compressor
Limited Range
Compressor,
Peak Limiter
Expandor
Noise Reducer
Volume
Vogad, Range
Reducer
Volume Lim-
iter, Half
Vogad
Range Restorer
classification which tells what the device is designed to do. In this
system the numbers preceding the letters specify the input control
range in decibels, and the position of a horizontal bar indicates the
position of the main signal range with respect to the control range.
The letters specify the time actions and in the case of vogads, where
several time actions may be combined, an arbitrary combination of
letters would be used. The final numbers specify the control ratio, and
in the case of vogads, where this might be different depending on
whether the input was increasing or decreasing, both values are given,
the former first.
In this system, definitions of time actions are prerequisite and by
way of illustration, the following symbols have been used:
/ represents instantaneous, meaning very fast adjustment of device
5' represents syllabic, meaning moderate speed adjustment of device
V represents volume, meaning a combination of controllers which
produces adjustment of device in response to dynamic speech
so that the output volume is approximately determined by the
input volume.
Secondary Characteristics
In addition to their three primary characteristics, range controllers
may have a number of secondary features which are sometimes im-
portant. The outstanding ones are:
1. Bias
A neutral range controller is one which holds its setting during the
quiet periods between words and sentences and which changes its gain
AMPLITUDE RANGE CONTROL
537
TABLE II
Comparison of Nomenxlature for Range Controllers
Col. (1)
Arbitrary ^
1. Vogad
2. Vogad Combined with
Syllabic Compressor
3. Volume Limiter
4. Compandor
5. Noise Reducer
6. Limited Range Com-
pressor
7. Peak Limiter
8. Peak Chopper
9. Crosstalk Suppressor
10. Rooter and Inverse
Rooter
n. Vodas (Singing Sup-
pressor Relay)
12. SvUabic Vodas
Col. (2)_
Systematic
Full Range 45 db
Volume Compressor
Full Range 45 db
Volume Compressor
High Range 15 db 1 : 5
Volume Compressor
Full Range 60 db 2 : 1
Syllabic Compandor
Low Range 20 db 2 : 1
Syllabic Expander
High Range 10 db 1 : 2
Instantaneous Compressor
High Range 12 db 1 : 5
Syllabic Compressor
Hi^h Range 6 db 1 : 100
Instantaneous Compressor
Low Range 10 db 10 : 1
Instantaneous Expandor
Full Range 70 db 2 : 1
Instantaneous Compandor
Col. (3)
Symbolic
45 F5/ 23-18
Compressor
45 F55/ 23-18
Compressor
15 VS Compressor
6052 Compandor
2052 Expandor
1072 Compressor
1255 Compressor
6/100 Compressor
10710 Expandor
7072 Compandor
07 so Expandor
05=0 Expandor
only when the waves acting on it differ from those just received. This
condition sets a new requirement on the range controller which can
usually be met by a combination of control circuits.
A biased controller is one which returns to a setting corresponding
to some fixed or biased intensity when speech is not passing and adjusts
itself each time speech begins. A simple compressor is biased since
with no input it generally takes a setting of maximum gain so as to be
right for the weakest waves that might be applied in its working range
or below the working range. It is also possible to bias a range control-
ler so as to have minimum gain, or any other intermediate value when
no waves are applied. An important secondary characteristic is the
rate at which the device returns to the desired "bias" point.
Any of the devices listed in the tables might be neutral or biased in
either direction, thus increasing the number of possible arrangements.
2. Behavior Outside of Range
For inputs outside the working range of a range controller it is
important to provide that the amplification of these waves does not
cause them to be modified so as to be out of proportion to output
signals in the main range. In some cases this is met by choosing a
device which follows the same law all the way to zero current. In
others, the device may act as a linear transducer, i.e., with range
538 BELL SYSTEM TECHNICAL JOURNAL
factor of one outside the working range. Various other combinations
of control ratios can, of course, be employed.
3. Pilot Channel
In all complete compressors some form of pilot channel is necessary
to control the re-expansion if this is required. If the gain changes
are slow, the pilot channel may include an operator who changes the
gain of the receiving device in a manner complementary to that of the
sending device based on aural or visual signals. If the gain changes
are too rapid for the operator to follow, the receiving gain may be
changed automatically.
The pilot channel itself may be a direct or alternating current of
variable amplitude or frequency, or in case of carrier or radio, it may
be the carrier frequency. In sound reproduction a pilot channel
might be a pilot track on the record.
Summary
In an amplitude range control system, the following characteristics
must be specified in addition to the usual repeater characteristics, to
determine its design and performance:
1. The steady-state control ratio, which determines how much control
is obtained and whether restoration can be made automatically
or not.
2. The manner in which the output varies with time, following a given
change in input.
3. The range over which it is to function.
In specific cases the following should also be considered:
4. The action of the device for inputs outside the working range.
5. If the device is a complete compressor, the type of pilot channel
for restoring.
6. The action of the device when signals are removed.
Acknowledgment
The computations used to obtain Figs. 3 to 9, inclusive, were made
by Miss Marian Darville.
References
1. " Devices for Controlling Amplitude Characteristics of Telephonic Signals," A. C.
Norwine. Presented at A. I. E. E. Pacific Coast Convention, Aug. 9-12, 1938.
2. "Speech Power and Its Measurement," L. J. Sivian, B. S. T. J., Vol. VIII,
pp. 646-661.
3. "The Compandor — An Aid Against Static in Radio Telephony," R. C. Mathes
and S. B. Wright, B. S. T. J., Vol. XIII, pp. 315-332.
4. "A Vogad for Radio Telephone Circuits," S. B. Wright, S. Doha, A. C. Dickieson.
Presented at I. R. E. Convention, June 1938.
Devices for Controlling Amplitude Characteristics of
Telephonic Signals*
By A. C. NORWINE
This paper describes a family of devices which automatically
respond to signals and control the circuit amplification in such a
way as to improve transmission. Their general characteristics are
outlined, their differences explained, and some of their applications
are listed.
Introduction
THE transmission of speech energy over electrical circuits is
attended by the interesting and sometimes difficult problem of
preserving the original signal in spite of limitations in the transmission
medium. These limitations include load carrying capacity, inter-
ference with other service, noise, change in attenuation with time and
many others. Because of special limitations it is sometimes desirable
to alter the amplitude characteristics of the speech or other signal
energy without, of course, materially lowering its intelligibility. In
high quality systems the peak voltage from some speech sounds of a
given talker may be over 30 db (some 30 times) higher than from his
weakest sounds when there is very little inflection in the speech.
Loudness changes for emphasis will increase this range of intensities.
Ordinary message systems do not have to contend with quite so wide a
range of instantaneous voltages from a single talker, but different
talkers under extreme terminal conditions produce about a 45 db range
of average voltage, which is additive to that for a single talker. Conse-
quently, a voltage range of about 70 db (over 3000 to 1) must be
considered for message circuits.
In order to accommodate such ranges of intensity to certain trans-
mission media such as radio links a new family of automatic devices has
been developed. In general all of these contain amplifiers or attenu-
ating networks whose loss or gain is changed according to some
function of the applied input and which may have a variety of time
sequences in their control circuits. It is hoped that by the classi-
fication and description of some of these devices their distinguishing
characteristics and fields of usefulness will be made somewhat clearer.
* Presented at the Pacific Coast Convention of A. I. E. E. and I. R. E. in Port-
land, Oregon, August 9-12, 1938.
539
540 BELL SYSTEM TECHNICAL JOURNAL
We are to be concerned here principally with those elements allied to
the telephonic art, although some applications are to be found in other
fields. It is not intended to include those voice operated functions
which are essentially switching operations although the distinction in
some cases becomes exceedingly fine.
Names of volume controlled devices * which have been used in
published papers^include vogad,t' ^' ^' * compandor, |' *• ^ and volume
limiter.^ Without direct comparison it may not be obvious how these
and similar devices differ. First the apparent similarity of several of
these devices will be shown in simple diagrams. Next the more
important characteristics of a number of devices will be presented in
tabular form, followed by descriptions of the different types. These
will then be discussed with particular emphasis on their distinctive
qualities, with notes on their variants which have some apparent
value.
General Characteristics of Volume Controlled Devices
In Figs. 1 to 10 are shown simplified diagrams of some of these
devices. While detailed descriptions of them will be deferred till later
it may be pointed out that all those shown contain vario-lossers, and all
have paths from the main transmission path to control circuits which
affect the vario-lossers. A vario-losser usually consists of a balanced
pair of vacuum tubes whose gain is changed by varying the grid bias, or
of a network of non-linear elements such as copper oxide or silicon
carbide whose loss is changed by varying a current through them. In
some special cases it may be a mechanically adjusted variable network.
The word vario-losser is thus a generic term relating to a circuit whose
loss or gain is controllable. A control circuit ordinarily consists of an
amplifier and rectifier whose direct current or alternating current
output bears a chosen relation to its input. Thus some control
circuits are marginal; they produce no control voltage till the input
exceeds some critical value, then produce large control voltages for
small additional increments of input. These are used, for example,
when it is desired to limit the output of a vario-losser to a definite
amount. Another type of control circuit produces a current or voltage
which is linear with input expressed in decibels. In combination with
a vario-losser whose gain is a linear function of control current or
voltage one can produce a device whose gain is a linear function in
decibels of the input to the control circuit.
* See the footnote on page 543.
t " Folume Operated Gain Adjusting Device."
X A combination of the names "Compressor" and "Expatidor."
AMPLITUDE CHARACTERISTICS OF TELEPHONIC SIGNALS 541
BIASED
CONTROL
CIRCUIT
VARIO-
LOSSER
GAIN
INCREASER
GAIN
DECREASER
BIASED
CONTROL
CIRCUIT
GAIN INCREASE DIS"
ABLER (BLOCKS
INCREASER ACTION)
VOGAD
BIASED
CONTROL
CIRCUIT
VARIO-
LOSSER
M
GAIN
INCREASER
A
GAIN DE-<
CREASER
GAIN INCREASE
DISABLER
VARIO-
LOSSER
BIASED
CONTROL
CIRCUIT
TIME FUNCTIONS SAME
AS VOGAD
GAIN CHANGES HALF
AS GREAT
HALF VOGAD
IN
VARIO-
OUT
,
BIASED 1
CIRC
:uiT
VOLUME LIMITER
FIGURE 3
REGULATING
THRESHOLD
IN
VARIO-
OUT
LOSSER
'
CONTROL
CIRCUIT
V
OLTAGE =
K- INPUT VULIAfct
COMPRESSOR
/ \
N
VARIO-
01.
LOSSER
.
CONTROL
CIRC
;uiT
K-
VOLTAC
INPUT \
;e =
/OLTAGE
EXPANDOR
JZI
542 BELL SYSTEM TECHNICAL JOURNAL
It will be recognized that if the application or removal of the control
energy is retarded, the action of the control circuit may be made quite
different on transient inputs than on steady state inputs. It will
appear later that this is the important distinction between some of the
devices to be discussed and that fundamental differences in their
functioning are thus brought about.
Referring to the figures once more it will be noted that some control
devices are connected to the transmission path at the input to the
vario-losser. These are known as "forward acting" control circuits.
Other controls, connected at the vario-losser outputs, are known as
"backward acting" control circuits. This is simply convenient
terminology to indicate whether the control energy is progressing in the
same direction as the main transmission or is progressing in a backward
direction after traversing the main path, usually through a vario-
losser. Some backward acting controls function to measure the
output of the devices containing them and to make whatever adjust-
ments are required. Others are placed in that position to take
advantage of the vario-lossers in the transmission paths, i.e., such
controls could be replaced by combinations of forward acting controls
and extra vario-lossers.
In Table I, nine of the volume controlled devices * which have been
developed for various commercial and experimental uses are listed with
the functions of voltage, time, and frequency which are employed to
obtain their respective performances. There is, of course, some
latitude in the choice of these functions for any one device. Pending
more complete description of the different types in the following
paragraphs this table should be viewed as illustrating the general
character of the different circuits and also the range of the variables
which already have been employed. For example, it will be seen that
instantaneous voltage of the signal wave, its short time average value,
peak power, syllabic variations, and long time average power have all
been used as criteria of gain settings in different circuits. Some
devices change their adjustments only when critical values or ranges
are exceeded, while others vary somewhat with every syllable if
speech, for example, is being transmitted. Some are linear transducers
to all but low or high amplitudes while others reduce or increase the
output range from that at the input. It will be seen that proper
choices of times for gain increase and gain decrease in combination
* The names employed do not follow an entirely logical classification, but they
are given here because they have had considerable usage. For the same reason the
term volume controlled devices is used, although to be strictly correct it might better
be sound energy controlled devices, for example, for not all the devices operate in
accordance with volume as measured by the well-known class of visual reading meters
called volume indicators.
AMPLITUDE CHARACTERISTICS OF TELEPHONIC SIGNALS 543
VARIO-
LOSSER
VOLTAGE =
K- INPUT VOLTAGE
CONTROL
CIRCUIT
LIMITED RANGE EXPANDOR
RADIO NOISE REDUCER
FIGURE 6
AMPLITUDE
A
AMPLITUDE
IN
VARIO-
OUT
LOSSER
;
CONTROL
1
VOLTAGE = ^
K-INPUT VOLTAGE
CIRC
:uii
IN
VARIO-
OUT
LOSSER
.
BIASED 1
CIRC
UIT
N
VARIO-
0
LOSSER
HIGHLY-
BIASED
CONTROL
CIRCUIT
AMPLI-
TUDE 8
PEAK CHOPPER
FIGURE 9
AMPLITUDE
BIASED
CONTROL
CIRCUIT
VARIO-
LOSSER
TWO VALUES
OF LOSS
CROSS-TALK SUPPRESSOR
FIGURE 10
AMPLITUDE
544
BELL SYSTEM TECHNICAL JOURNAL
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546 BELL SYSTEM TECHNICAL JOURNAL
with certain gain control criteria make possible a wide variety of signal
altering means to meet different requirements.
Description of Devices in Table I
With this introduction to the combinations of characteristics which
are possible it should be less difficult to distinguish between the specific
devices discussed in the following paragraphs, which, in addition to
describing the devices, contain some comments which should assist in
visualizing their forms and their operation.
1. The vogad (Fig. 1) is a device which will maintain at its output
speech volume ^ which, over a certain range of input, is relatively
independent of the speech volume applied to its input and which, in
the ideal case, will not change its gain during periods of no speech input.
It makes little or no alteration in the ratios of maximum and minimum
instantaneous to average voltages of the speech.
2. The volume limiter (Fig. 3) is a device which is a linear transducer
for all speech volumes up to a critical value, beyond which all input
volumes produce essentially the same output volume. It is essentially
different from the vogad in that its gain approaches the maximum
value when input is removed.
3. The compandor (Figs. 4 and 5) is composed of a compressor and an
expandor. A compressor is a device whose input-output characteristic
on a decibel scale has a slope less than unity * and whose gain or loss is
variable under control of the input energy at a time rate which will
permit it to follow the syllabic rate of change of speech energy. Simi-
larly, an expandor is a device whose input-output curve has a slope
greater than unity and whose gain is variable at a syllabic rate under
control of the input energy. Thus very shortly after all input is
removed the gain of a compressor is maximum and the loss of an
expandor is maximum. The reciprocal of the compressor characteristic
slope is spoken of as the compression ratio, and the slope of the ex-
pandor characteristic is spoken of as the expansion ratio.
4. The radio noise reducer *• ^ (Fig. 6) combines the functions of an
expandor which operates in the range of amplitudes where noise and
weaker speech sounds lie and a linear transducer which comes into play
for all amplitudes exceeding a critical value, which can be set to best
suit the atmospheric noise conditions. In other words, the radio noise
reducer is a limited range expandor. Inputs which are below the
expandor range are subject to transmission at the minimum gain.
5. The limited range compressor (Fig. 7) is a device whose operating
* That is, if the input increases by x db the output increases by less than x db.
AMPLITUDE CHARACTERISTICS OF TELEPHONIC SIGNALS 547
range includes a region within which compression at a syllabic rate can
take place; at other inputs the device is a linear transducer. Its
connecting diagram and time functions are the same as those shown in
Fig. 5 except that the control circuit contains a limiting device, so that
compression takes place in only a portion of its input range, analogous
to the action of the limited range expandor of Fig. 6. As a special case
the limited range compressor may have no linear range above its
compression range, thus becoming one type of peak limiter.
6. The peak limiter (Fig. 8) is a device whose gain will be quickly
reduced and slowly restored when the instantaneous peak power of the
input exceeds a predetermined value. The amount of gain reduction is
a function of the peak amplitude, and in practice is usually intended to
be small to prevent material reduction of the range of intensity of the
signal.
7. The peak chopper (Fig. 9) is a device which prevents transmission
of peak amplitudes exceeding a critical amount, an essential charac-
teristic being that the loss it inserts is completely determined by the
instantaneous voltage of the signal. That is, its operating and
releasing times are substantially equal to zero.
8. The crosstalk suppressor (Fig. 10) is a device which normally
presents a prescribed loss to transmission, which loss is removed
rapidly when the input amplitude exceeds a certain threshold and is
reinserted at a definite time after the input is removed. It reduces low
amplitude unwanted currents such as crosstalk but does not affect
amplitudes in the useful signal voltage range. This device differs from
the limited range expandor in that the time during which the low loss
condition is maintained is considerably greater, so the transition from
one gain to the other occurs less frequently.
9. A rooter is an instantaneous compressor. Such a circuit can be
made to produce an output whose instantaneous voltage is, for example,
the square root or some similar function of the instantaneous voltage
applied to the input. An inverse rooter is an instantaneous expandor
whose characteristic is complementary to that of the rooter. A
combination of rooter and inverse rooter will reduce the load require-
ments on a transmission system between the two units but requires
that it transmit a wider band of frequencies than that for the original
signal, and that it be essentially free from phase distortion. This does
not seem to be an attractive arrangement from a commercial viewpoint
and is included here simply as an illustration of one of the possible
modifications of signal energy. It is not shown" in the group of
diagrams.
548 BELL SYSTEM TECHNICAL JOURNAL
Variants to the Devices Described
In addition to these there are various devices which are essentially
modifications of those described. For example, a half-vogad, Fig. 2,
may have the same time functions as a vogad, Fig. 1, but the gain
changes in the transmission circuit are half as great for the same range
of input volumes. Thus in a vogad the range of gain changes in the
transmission circuit is equal to the range of input volumes, so that the
output volume is the same for all input volumes. In the case of the
half vogad the range of gain changes in the transmission path is one-
half the range of input volumes, so the output volume range is one-half
that of the input. It is also possible to construct a vogad whose output
volume range is any desired fraction of the input range. As another
example of modification of the devices described, for special appli-
cations it may be desirable to incorporate a certain amount of syllabic
compression in a vogad.
Communication circuits which have separate paths for oppositely
directed transmission between the two terminals are usually operated
at such an overall loss that with ordinary terminations there will be
little tendency for circulating currents to build up to a "singing"
condition. Sometimes there may not be a great deal of margin,
however, so that volume controlled devices added to such circuits must
add loss at some point to counterbalance whatever gain is put in at
some other point. Thus a vogad inserted at the transmitting side of
one terminal of such a circuit to amplify speech energy from weak
talkers must be supplemented by a "reverse vogad" in the receiving
side of the circuit. The reverse vogad is simply another vario-losser
which is operated upon by the vogad control circuit in such a way that
it always has a loss numerically equal to the gain of the vogad. Any
vogad gain will be compensated by the reverse vogad loss, so no
greater tendency to sing will be effected by the addition of the combi-
nation to the circuit. In like manner half vogads must be used with
compensating reverse half vogads.
Combinations of some of the devices also have interesting charac-
teristics. For example, a combined radio noise reducer and peak
limiter at the receiving end of a circuit would suppress noise and would
also reduce the amplitude of excessively high amplitude signals.
Likewise, a vogad, compressor, and peak chopper in tandem in the
order named could be made to reduce the range of input signals by a
very large amount for transmission over a medium having only a small
range between noise and maximum permissible signal. In this case it
would be practically impossible to recover the original signal range at
AMPLITUDE CHARACTERISTICS OF TELEPHONIC SIGNALS 549
the receiving terminal of the medium, but the intelligibility of speech
over such a system has been shown in the laboratory to be good.
Special compandors for high quality service may require compression
and expansion which vary with frequency. The exact characteristics
will depend upon band width, program material and transmission
medium. For transmission media in which the noise reproduced at the
receiving end is principally at the higher frequencies an unusual effect
is obtained if the usual variety of compandor is used. Low frequencies
unaccompanied by high frequencies will cause a gain change in com-
pressor and expandor, thus changing the background of high-frequency
noise which is not masked by the low-frequency signal energy. The
resulting swishing noise has been given the onomatopoeic name of
"hush-hush effect." To avoid this, recourse may be had to split band
compandors in which the compression and expansion is done only at
high frequencies or separately for low and high frequencies. The
successful application of the latter method is, however, more difficult
than it appears from its simple description.
Distinguishing Characteristics
It is important to distinguish between the half vogad. Fig. 2, and
the compressor. Fig. 4. As shown in Table I the latter operates on
syllabic variations and the former on the average volume of the input.
Thus the half vogad reduces the range of output volumes to one-half
that at the input while the compressor reduces the range of syllabic
power at its output to one-half that at the input. In other words, the
compressor reduces the ratio of peak to average power on constant
volume speech, while the half- vogad simply adjusts for that volume and
does not alter the peak ratio. There is, of course, the additional
important difference that the half-vogad retains its gain setting during
silent periods while the compressor, by virtue of having followed the
syllabic power, has its maximum gain during silent periods.
Volume limiters. Fig. 3, may be mistaken for vogads. Fig. 1, because
during speech input above a certain value the two may produce the
same output volume. They both employ something like a measure-
ment of average power over periods longer than a syllable to determine
their gain settings. The important difference is that a vogad retains
its gain setting when speech currents are not present, while a volume
limiter approaches its maximum gain during such periods. In terms of
the output resulting from a range of input volumes there is another
important difference if the volume limiter operates over only part of the
input range: the vogad reduces the width of the distribution curve of
volumes to a very small value, while the volume limiter moves all the
550
BELL SYSTEM TECHNICAL JOURNAL
area under the distribution curve above a certain point to the region
near that point, which is its Hmiting volume. This is illustrated in
Fig. 11, in which the calculated modifications of a volume distribution
by a vogad and by a volume limiter are shown. In the cases "without
volume control " and "with a vogad " the distributions are normal, and
the standard deviation, a, has its usual statistical significance. With a
volume limiter, only volumes above the limiting volume are affected,
WITH vogad/
(cr = I DB) 1
1 c
/ /
1 /
//
//
//
WITHOUT //
VOLUME CONTROL //
) 1
\ 1
\ 1
\ \
\ \
\ \
\\ +4
V
(Cr=l DB FOR VOLUMES
ABOVE 0,-1-4, AND +7 DB)
\ +7
(0-/ L)B2__^,— -*^ — 7
■ 1 r y
n
-15 -10 -5 0 5 10 15
VOLUME IN DECIBELS FROM MEAN VOLUME
Fig. 11 — Modification of volume distribution by use of a vogad or a volume limiter.
and these higher volumes are redistributed according to a normal law
whose standard deviation is 1 decibel, as stated in the figure.
It is also important to distinguish between a peak limiter and a peak
chopper. Figs. 8 and 9. Naturally they resemble one another since
they are intended to permit transmission of signals at higher average
amplitudes without excessive loading of transmission circuits. How-
ever, they are intended for different classes of service and hence are not
interchangeable except in some borderline cases. For the highest
grade of transmission harmonic production must be negligible and the
reduction in amplitude range of signals small and infrequent. Gain
changes must be smooth, though rapid enough to compensate for
practically any input wave to be expected. These characteristics are
found in the peak limiter now being furnished for use on program
networks and radio transmitters."- ^^ For services in which it is
desirable to maintain the signal energy at a high value to over-ride
noise and in which harmonic distortion must be kept low a peak
limiter with somewhat smaller time constants may be used. A high
ratio limited range compressor might be suitable in this instance.
This device would lower its gain a little more quickly on excessive
AMPLITUDE CHARACTERISTICS OF TELEPHONIC SIGNALS 551
inputs, and it would also reinsert its gain much more quickly; it would
affect the naturalness of the sound of the signal more than the slower
peak limiter but it would also cause the signal to over-ride noise some-
what better. In a third variety of service the harmonic distortion
introduced by a limiter is a secondary matter, the prime consideration
being that the peak amplitude of the signal shall not exceed a specified
value. This may be because higher amplitude signals would produce a
tremendous increase in distortion or crosstalk into other channels or
would damage expensive equipment farther along in the circuit. For
these cases we may use the fastest possible type of limiter, the peak
chopper, which simply cuts off any peak exceeding a certain value.
The crosstalk suppressor, Fig. 10, is a splendid example of the fine
distinction between volume controlled and voice operated switching
devices. This device has been described, but in the present state of
the art its time functions have not been definitely fixed. If the
characteristic of loss versus input is made steep enough and the speed
of operation fast enough it will sound like a switching circuit and may in
fact be replaced by a relay-switched attenuating network. If made
somewhat slower and given a smaller slope of loss versus input it
approaches the limited range expandor or noise reducer.
Applications and Expected Advantages
It may be of interest to give some approximate figures on the
magnitudes of the advantages to be obtained by the use of some of
these devices. It will be understood that the values to be given are
simply illustrative, some having been obtained from field service on
particular models and some from tests on laboratory equipment under
special conditions.
Vogads appear to be most useful in such circuits as transoceanic
radio connections, where it is important to properly operate the
terminal switching equipment and to transmit over the radio circuit
speech energy from loud and weak talkers equally well. It is essential
in such cases that noise should not be increased in amplitude during
speech pauses, hence the gain retaining feature of the vogad. On such
a circuit a vogad will reduce a 45 db volume range to about 2 to 4 db.
This is equivalent to expert manual volume control.
Volume limiters are in use at the present time to prevent peaks of
speech energy in carrier circuits from "splashing" into telegraph
channels. '^ Some 5 to 10 db limiting is allowed on loudest talkers,
which causes little degradation of the speech channels but makes
possible the use of telegraph on the same carrier system. There is no
552 BELL SYSTEM TECHNICAL JOURNAL
wide-spread use of volume limiters in point-to-point radio service so
far, but in cases in which there is no disadvantage in raising noise in
silent periods in speech, such as in push-to-talk installations, proper
transmitter loading can be obtained with volume limiters fairly
cheaply.
One commercial model peak limiter, used as part of a program
amplifier ^^' " is capable of introducing a considerable amount of
compression without overloading on peaks, but for the preservation of
adequate program volume range it is being recommended that only
3 db peak limiting be allowed. This, of course, reduces the range of
intensity of the program, but from the standpoint of the listeners it is
equivalent to doubling the transmitted power or obtaining the same
signal-to-noise ratio with half the transmitted power.
Limited range compressors might be used either on land lines to
insure full loading or on radio links whose fading is too severe to permit
the use of normal compandors. There is no commercial application of
either sort at the present time. Peak choppers are, however, used on
some high power radio transmitters which might otherwise be tempo-
rarily disabled by high peaks in the signal being transmitted.
The chief usefulness of compandors is on radio links in which the
transmission of a compressed signal with subsequent expansion permits
operation through higher noise or with lower transmitter power. On a
long-wave transatlantic radio telephone circuit a compandor with
40 db range has been shown to allow an increase in noise of some 5 db
before reaching the commercial limit.^ With smaller amounts of noise
the noise advantage of the compandor approaches half its range in
decibels. This benefit is sometimes applied to a reduction of trans-
mitter power.
Radio noise reducers have been used to advantage in connection
with short-wave ship-to-shore and transoceanic radio telephone
service. In the former, routine transmission rating is given on a
judgment basis using a merit scale from 1 to 5, 5 being practically
perfect transmission and 1 so poor that intelligibility is very close to
zero. It will then be seen that the observed improvement of J^ to 1
point in transmission rating due to the noise reducer is of considerable
importance. Perhaps more graphic figures are those for transoceanic
service, where the reduction of noise in the receiving path not only
reduces the noise heard by the listener but also improves the voice
operated switching with the indirect result that at times .receiving
volume increases of 5 to 15 db are realized.^
As has been noted, the radio noise reducer is a special use of an
expandor alone. There are also two interesting applications for a
AMPLITUDE CHARACTERISTICS OF TELEPHONIC SIGNALS 553
compressor alone. The first, which uses a fairly high ratio of com-
pression, has been mentioned as one type of peak limiting device. The
second, using a moderate ratio of compression, is in connection with
announcing systems for use in very noisy locations. Its effect is to
amplify weak sounds more than strong sounds, which considerably
improves the intelligibility through high noise. For quiet locations it
is of less value, since the speech sounds lose some of their naturalness in
this process.
Conclusion
In the course of developing various types of the volume controlled
devices which have been described means have been worked out for
providing almost any combination of time constants, range of control,
and other characteristics which may be required. Some devices for
which there were specific commercial applications or useful functional
characteristics for experimental work have been constructed, with
resulting advantages which have been briefly mentioned. There
remain many possible ways to alter the characteristics of signal energy
such as speech to which these methods are applicable and which await
the special needs of new transmission problems.
Bibliography
1. C. C. I. F. White Book, 1 bis, pp. 77, 343.
2. C. C. I. F. White Book, 1 bis, pp. 251-3.
3. "A Vogad for Radio Telephone Control Terminals," S. Doba, Jr., Bell Labora-
tories Record, Oct. 1938, Vol. 17, No. 2, pp. 49-52.
4. "A Vogad for Radio Telephone Circuits," S. B. Wright, S. Doba, Jr., and A. C.
Dickieson, Presented at /. R. E. Convention in New York, June 18, 1938; to
be published in Proc. I. R. E.
5 "The 'Compandor' — An Aid Against Static in Radio Telephony," R. C. Mathes
and S. B. Wright, Elec. Engg., 1934, Vol. 53, No. 6, pp. 860-6; Bell Sys. Tech.
Jour., July 1934, Vol. 13, No. 3, pp. 315-32.
6. "The Voice Operated Compandor," N. C. Norman, Com. and Br. Engg., Nov.
1934, Vol. 1, No. 1, pp. 7-9; Bell Lab. Record, Dec. 1934, Vol. 13, No. 4,
pp. 98-103.
7. "Volume Limiter Circuits," G. W. Cowley, Bell Lab. Record, June 1937, Vol. 15,
No. 10, pp. 311-15.
8. "A Noise Reducer for Radio Telephone Circuits," N. C. Norman, Bell Lab.
Record, May 1937, Vol. 15, No. 9, pp. 702-7.
9. "Radio Telephone Noise Reduction by Voice Control at Receiver," C. C.
Taylor, Elec. Engg., Aug. 1937, Vol. 56, No. 8, pp. 971-4, 1011; Bell Sys.
Tech. Jour., Oct. 1937, Vol. 16, No. 4, pp. 475-86.
10. "Higher Volumes Without Overloading," S. Doba, Jr., Bell Lab. Record, Jan.
1938, Vol. 16, No. 5, pp. 174-8.
11. "A Volume Limiting Amplifier," O. M. Hovgaard, Bell Lab. Record, Jan. 1938,
Vol. 16, No. 5, pp. 179-84.
For the sake of completeness the following references are included, although no
allusion has been made to them under the specific device-names used in this paper.
12. "tJber automatische Amplitudenbegrenzer," H. F. Mayer, E. N. T., 1928,
Vol. 5, No. 11, pp. 468-72.
554 BELL SYSTEM TECHNICAL JOURNAL
13. "High Quality Radio Broadcast Transmission and Reception," Stuart Ballan-
tine, Proc. I. R. E., May 1934, Vol. 22, No. 5, pp. 564-629.
14. "Expanding the Music," A. L. M. Sowerby, Wireless World, Aug. 24, 1934,
Vol. 35, No. 8, pp. 150-2.
15. "Extending Volume Range," Radio Engg., Nov. 1934, Vol. 14, No. 11, pp.
7-9, 13.
16. "Amplitudenabhangige Verstarker," W. Nestel, E. T. Z., 1934, Vol. 55, No. 36,
pp. 882-4.
17. "An Automatic Volume Expandor," W. N. Weeden, Electronics, June 1935,
Vol. 8, No. 6, pp. 184, 5.
18. "Die Arbeitsweise der selbsttatigen Regelapparaturen," H. Bartels and W. G.
Ulbricht, E. N. T., 1935, Vol. 12, No. 11, pp. 368-79.
19. "Practical Volume Expansion," C. M. Sinnett, Electronics, Nov. 1935, Vol. 8,
No. 11, pp. 428-30, 446.
20. "Light-bulb Volume Expandor," Electronics, Mar. 1936, Vol. 9, No. 3, p. 9.
21. "Simplified Volume Expansion," W. N. Weeden, Wireless World, Apr. 24, 1936,
Vol. 38, No. 17, pp. 407-8.
22. "Practical Volume Compression," L. B. Hallman, Jr., Electronics, June 1936,
Vol. 9, No. 6, pp. 15-17, 42.
23. "Notes on Contrast Expansion," Gerald Sayers, Wireless World, Sept. 18, 1936,
Vol. 39, No. 12, p. 313.
24. "Contrast Amplification: A New Development," W. N. Weeden, Wireless World,
Dec. 18, 1936, Vol. 39, No. 25, pp. 636-38.
25. "Overmodulation Control and Volume Compression with Variable-mu Speech
Amplifier," W. B. Plummer, Q. S. T., Oct. 1937, Vol. 21, No. 10, pp. 31-33.
26. "Limiting Amplifiers," John P. Taylor, Communications, Dec. 1937, Vol. 17,
No. 12, pp. 7-10, 39-40.
27. "Low Distortion Volume Expansion Using Negative Feedback," B. J. Stevens,
Wireless Engr., Mar. 1938, Vol. 15, No. 174, pp. 143-9.
28. "Distortion Limiter for Radio Receivers," M. L. Levy, Electronics, Mar. 1938,
Vol. 11, No. 3, p. 26.
29. "Automatic Modulation Control," L. C. Waller, Radio, Mar. 1938, No. 227,
pp. 21-6, 72, 74.
30. "An AVE Noise Silencer Unit," McMurdo Silver, Radio News, May 1938,
Vol. 20, No. 11, pp. 46, 55.
The Exponential Transmission Line *
By CHAS. R. BURROWS
The theory of the exponential transmission line is developed.
It is found to be a high pass, impedance transforming filter. The
cutoff frequency depends upon the rate of taper.
The deviation of the exponential line from an ideal impedance
transformer may be decreased by an order of magnitude by shunt-
ing the low impedance end with an inductance and inserting a
capacitance in series with the high impedance end. The magni-
tudes of these reactances are equal to the impedance level at their
respective ends of the line at the cutoff frequency.
For a two-to-one impedance transformer the line is 0.0551 wave-
lengths long at the cutoff frequency. For a four-to-one impedance
transformer the line is 0.1102 wave-lengths long at the cutoff
frequency, etc.
The results have been verified experimentally. Practical lines
50 meters and 15 meters long have been constructed which trans-
form from 600 to 300 ohms over the frequency range from 4 to 30
mc. with deviations from the ideal that are small compared with
the deviations from the ideal of commercial transmission lines,
either two-wire or concentric.
When an exponential line is used as a dissipative load of known
impedance instead of a uniform line it is possible to approach more
nearly the ideal of constant heat dissipation per unit length.
This makes it possible to use a shorter line.
THE exponential line may be defined as an ordinary transmission
line in which the spacing between the conductors (or conductor
size) is not constant but varies in such a way that the distributed
inductance and capacitance vary exponentially with the distance along
the line. That is, the impedance ratio for two points a fixed distance
apart is independent of the position of these two points along the line.
A disturbance is propagated down an exponential transmission line in
the same manner as it would be down a uniform line with the addi-
tional effect that the voltage is increased by the square root of the
change in impedance level and the current is decreased by the reciprocal
of this quantity.
The exponential line has the properties of a high pass impedance
transforming filter. The cutoff frequency depends upon the rate of
* Presented before joint meeting of U. R. S. I., and I. R. E., Washington, D. C,
April 1938. Published in Communications , October 1938.
555
556 BELL SYSTEM TECHNICAL JOURNAL
taper. As the frequency is increased the transfer constant * ap-
proaches the propagation constant of the equivalent uniform Hne. At
sufficiently low frequencies the only effect of the line is to connect the
input to the load.
Above cutoff the magnitudes of the characteristic impedances at any
point are approximately equal to the nominal characteristic imped-
ance * at that point but their phase angles (in radians) differ by an
amount which at the higher frequencies is equal to the cutoff frequency
divided by the frequency in question. The ratio of input impedance
to the input impedance level * of an exponential line terminated in a
resistance equal to the impedance level at the output always remains
within the range from 1 — fi/fto 1/(1 — fi/f) for frequencies,/, greater
than the cutoff frequency, /i. For a 2 : 1 transformation this means
that the input impedance remains within ± 6 per cent of the desired
value for all frequencies above that for which the line is a wave-length
long. For a 4 : 1 transformation under the same conditions the
irregularities are twice as great.
A transforming network having deviations from the ideal of the
order of ± (fi/fY may be made by connecting an inductance in parallel
with the low impedance terminal and a capacitance in series with the
high impedance terminal. The magnitudes of these reactances are such
that their impedances are equal to the impedance levels of the line at
their respective ends at the cutoff frequency. Or expressed in another
way the capacitance is equal to 2/(k — 1) times the electrostatic
capacitance of the line and the inductance is the same factor times the
total loop inductance of the line where k is the impedance transforma-
tion ratio of the line.
Figure 1 shows the theoretical input impedance-frequency charac-
teristics for 2 to 1 step-up and step-down exponential lines. Curve 1
is for the line with a resistance termination. At low frequencies the
input impedance is equal to the load impedance while at high fre-
quencies the line approaches an ideal transformer. Curve 2 is the
input impedance of the line terminated with the appropriate resistance-
reactance combination. The improvement in the input impedance
characteristic for frequencies above the cutoff frequency is evident.
At the lower frequencies the input impedance does not approach the
terminal reactance but approaches the reactance of the capacitance of
the line in parallel with the series terminal capacitance for the step-up
line and the reactance of the inductance of the line in series with the
shunt terminal inductance for the step-down line. The improvement
is not as great as apparent from the figures because the phase angle is
* See appendix for definition of terms.
EXPONENTIAL TRANSMISSION LINE
557
not improved proportionally. This is easily remedied by completing
the impedance transforming network with the appropriate reactance
at the input. The resulting input impedance is shown in curve 3. In
the "pass" frequency range the maximum reactive component is of
0.2
as
1.0
RELATIVE FREQUENCY f/f
2.0 5.0 •
10
20
50
N. '
-T T -T-r T 1 ' 1 1 111 '
1 1
' '
'
•
• '
' '
~x —
1 III II
i. X
CUT OFF frequ:ncy
VX
f,= 0.0551 fn
5.0
\
s.
1
Kline
IS l/8 A LONG
''-LINE IS ONE
WAVELENGTH
LONG AT THIS
FREQUENCY
^
N
AT THIS FREQUENCY
^
^v
S
s
\
^
V
"V,
--
g -TERMINAL IMPEDANCE CASE 2 8. CASE 3
-
1.0
■^
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s
^ . Verm
INAL
IMP
EDA
N(
:e
c
,A<
>E 1
-
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^
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y
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r
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/
\
//
N
//
/
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.
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o.a
/
:
0.1
0.5 ti
2.0 i
I
3.0 :
001 0-05 0.1 0-5 1.0 4"
RELATIVE FREQUENCY f/fQ
Fig. 1 — Input impedance characteristics of 1 : 2 exponential lines. Left ordinate
scale refers to step-up line. Right ordinate scale refers to step-down line.
Curve 1— Resistance termination.
Curve 2 — With capacity equal to twice the electrostatic capacity of the line in series
with the same resistance, Zj = Zi.{\ — jfilf), for step-up line, or with an in-
ductance equal to twice the total inductance of the line in shunt with the
same resistance, Zi = Zi/(1 — jfi/f), for step-down line.
Curve 3 — Termination as for curve 2 with inductance equal to twice the total induc-
tance of the line in parallel with input to the line, Zn = Zi/{1 — jfi/f), for
step-up line, or termination as for curve 2 with capacity equal to twice the
static capacity of line in series with input to the line, Za = Zi{\ — jfi/f).
Curve 4 — Asymptotic value of impedance of capacity of line in parallel with termina-
tion for case 2 for step-up line, or asymptotic value of impedance of inductance
in series with termination for case 2 for step-down line.
Curve 5 — Impedance of shunt inductance added at input for case 3 for step-up line,
or impedance of capacity added in series at input for case 3 for step-down line.
the same order of magnitude as the deviation of the impedance from
the ideal.
Besides its application as an impedance transforming network, the
exponential line may be used as a "resistance" load of constant known
impedance that has a high capability for dissipating power. As such
it is capable of dissipating more power in the same length of line than
558
BELL SYSTEM TECHNICAL JOURNAL
the uniform line. If x is the maximum attenuation in nepers that can
be obtained with a uniform line without overheating, the same length
of exponential line will have an attenuation of (e^^ — l)/2 nepers.
Exponential lines of the proper length have properties similar to
half-wave and quarter-wave uniform lines. The input impedance of
an exponential line an even number of quarter wave-lengths long is
equal to the load impedance times the impedance transformation ratio
of the line. When the length of the line differs from an odd multiple
of a quarter wave-length by an amount that depends upon the fre-
quency and load impedance, the input impedance is equal to the product
of the terminal impedance levels divided by the load impedance.
Mathematical Formulation
The telegraph equations for the exponential line may be solved by
the methods employed in the problem of a uniform line. The resulting
equations for the voltage and current at any point along the line are
and
tx
Vx = Ae
5]
-(-|)^a.R+(^+l>-..-(^-|)
+ Be
1 + I e^^^
-4^ 2-(r+|> ^^ + 2;-i-(r-|>
Zo 7
Za 7
(1)
where
Zq 7
1 -
B^ + 2
r -
(2)
^ ^ log^o ^ log^ ^ log^o .^ ^^^ ^^^^ ^f ^^p^^^
Zj; = Vz/y — Z^e^"" is the surge or nominal characteristic impedance of
the exponential line at the point x which is equal to the
characteristic impedance of the uniform line that has the
same distributed constants as this line has at the point x,
Y = "^zy = Vzo3'o is the propagation constant of any uniform line that
has the same distributed constants as this line at any
point. It is independent of the point along the line to
which it is referred, and
r = V7^ + 5-/4 = a + j(8 is the transfer constant of the exponential
line.
EXPONENTIAL TRANSMISSION LINE 559
+ 7 and + r refer to the values of the indicated roots that are in the
first quadrant.
If these equations are compared with those for a uniform trans-
mission Hne it is found that the propagation constant is F — 5/2 for
voltage waves traveling in the positive x direction and F + 5/2 for
voltage waves traveling in the negative x direction. For current
waves the corresponding propagation constants are F + 5/2 and F — 5/2.
In the terminology of wave filters, F is the transfer constant and 5 is
the impedance transformation constant. 5/2 is the voltage transformation
constant and — 5/2 is the current transformation constant. The real
and imaginary parts of F, a and /3 are the attenuation and phase
constants respectively.
An important parameter is
. 5
which for a non-dissipative line is the ratio of the cutoff frequency to
the frequency, as can be seen if we write the transfer constant as
r = 7 Vi - v"^,
where the indicated root is in the fourth quadrant. For a non-
dissipative line V is real and the transfer constant is real or imaginary
depending on whether v^ is greater than or less than unity. Hence the
exponential line is a high pass filter whose cutoff frequency, /i, is that
frequency for which j^ = ± 1. The transfer constant is then less than
that for a uniform line by the factor Vl — i'^ so that both phase velocity
and wave-length are larger for the exponential line than for the uniform
line by the reciprocal of this factor.
If we terminate this line dit x — I with an impedance Zi = vi/ii, the
ratio of the reflected to direct voltage wave is found to be
A 1 + {z,izdNi - p' -jv) '
where the coefficient of the exponential is the voltage reflection coefficient.
There will be no reflection if
Zi = ZiK^TT^^- + jp) = z,+, (4)
which becomes Zie~'^^^~^ " above the cutoff frequency for non-
dissipative lines. This is the magnitude of the forward-looking
characteristic impedance at x = I as can be seen by dividing the first
term of (1) by the first term of (2). Curve 1 of Fig. 2 gives the charac-
560
BELL SYSTEM TECHNICAL JOURNAL
teristic impedance of a non-dissipative exponential line looking toward
the high impedance end as a function of frequency. At infinite fre-
quency the characteristic impedance is a resistance equal to the
nominal characteristic impedance but as the frequency is decreased the
phase angle of the characteristic impedance changes so that its locus
+J
V-
®
0.9
-\
®
■^
^
0.8
/
^N
»-i^
0^-°
0.7
0.5
f
T
>
0.2
0
(
J©
Sd
^^L
2.0/—
1.3
-J
1 ____--
%
-^
^
i
^^
0.9
1
V^i
iV^
Fig. 2 — Impedance diagram comparing the forward looking characteristic imped-
ance with various terminal impedances. The numbers give the frequency relative
to cutoff. The arrows are the vectors Zj — Zj"*" which are a measure of the magnitude
of the reflection.
A. Step-up line.
Curve 1 — Forward looking characteristic impedance,
Zi+ = Zie-^^'''~'^^^'f\ f>fi,
Zi^ = Zil- i(/i//)(l 4- y!\-Plim, /i > /;
Curve 2 — Resistance termination, Zi = Zi;
Curve 3 — Capacity resistance termination, Zi = Zi{\ — jf\lf);
Curve 4 — Capacity, resistance and inductance termination adjusted for no reflection
at twice the cutoff frequency and at infinite frequency;
B. Step-down line.
Curve 5 — Forward-looking characteristic impedance,
Zi+ = Z;[+i(/>//)(l - Vl -/V/i^)], /i > /;
Curve 6 — Resistance termination Z/ = Zj;
Curve 7 — Inductance resistance termination Z; = Z;/(l — jfijf);
Curve 8 — Inductance, resistance and capacity termination adjusted for no reflection
at twice the cutoff frequency and at infinite frequency.
EXPONENTIAL TRANSMISSION LINE 561
is the circular arc. At and below cutoff it is a pure reactance. If the
load is a resistance equal to the nominal characteristic impedance at
the terminal as indicated at 2 of Fig. 2, there will be no reflection at
infinite frequency, but as the frequency is lowered there will be an in-
creasing impedance mismatch with its accompanying reflected wave.
This reflection may be materially reduced by inserting a condenser
in series with the resistance load as shown by curve 3. Further im-
provement results from more complicated networks. Curve 4 shows
the effect of adding an inductance in shunt with the resistance load of
the resistance-capacitance combination. The arrows indicate the
resulting impedance mismatch which is a measure of the reflected wave.
The characteristic impedance looking toward the low impedance end
is the inverse of that looking in the other direction as shown by
curve 5. Shunting the resistance load with an mductance gives the
impedance curve 7. Adding a capacitance element gives curve 8.
Division of (1) by (2) and substitution of the result of (3) gives the
following ratio for the impedance looking into the line at the point x
to the impedance level at that point,
Z^ _ K{^\ - y-" -jv) -f 1 -f- \K{^\ - t'^ -f-jV) - l]g-^r(;-.)
Z- K + jv + Vl - v' - [K - Vr^T^ + jV>-2r('--)
where K = ZijZi is the ratio of the load impedance to the impedance
level at the terminal. Here as before the indicated root is in the
fourth quadrant.
Network Characteristics
Three parameters are required to specify the characteristics of an
exponential line of negligible loss: (1) the cutoff frequency, /i, (2) the
length of the line which is perhaps best specified as the frequency,
/o = velocity of light/length of line, for which the line is one wave-
length long, and (3) the impedance level at some point along the line.
We will designate the impedance levels at the low and high impedance
ends of the line by Zi and Zi respectively, and their ratio Z2/Z1 by k.
When the line is terminated in a resistance equal to the impedance
level at the output (5) reduces to
Zi , „. o , 1 + i tan f k-"""^ 2r
Zi 1 - 7 tan f k"''^ -f ' ^ ^
^1 ,_„, 1 +tan^/^^ + i-'')
^' H-tan^/-'^-2-'')
562
BELL SYSTEM TECHNICAL JOURNAL
for frequencies below and above cutoff respectively. Here r? = — j2Yl
is twice the electrical length of the line in radians, sin 2f = Ijv,
sin 2^ = V and cos 2^ is ratio of the electrical length of the line to that
of a uniform line of the same physical length. For the step-down line
the corresponding ratios are the reciprocal of the above expressions.
These ratios are plotted in Fig. 1.
When /— ^ 0, £i = kZi = Zi — Zi and the only effect of the line is
to connect the load to the input. Above cutoff the magnitude of the
input impedance oscillates about the nominal characteristic impedance
and the phase angle oscillates about the value — 2^(« — /i// for
/>J> /i) which goes from — 7r/2 to 0 as the frequency increases indefi-
0.1
0.5 1.0
FREQUENCY IN MEGACYCLES
Fig. 3 — Input impedance characteristics.
Curve 1 — 150 : 600 ohm line, 100 meters long.
Curve 2—300 : 600 ohm line, 200 meter? long.
Both lines have the same rate of taper.
nitely from cutoff. The variation of the input impedance with fre-
quency is shown for two lines of different length but the same rate of
taper in Fig. 3. The magnitude of the oscillations depends only on
the rate of taper and decreases with increase in frequency. The
impedance varies between (1 + /i//) and 1/(1 + /i//). The positions
of the maxima and minima, however, are determined by the length of
the line. They occur respectively at those frequencies for which the
line is approximately 1/8 of a wave-length more than an even or an odd
number of quarter wave-lengths long. The phase angle is usually
negative but has a small positive value when the line is approximately
a half wave-length long.
The locations of these maxima and minima are the same as would
result from terminating a uniform line in an impedance whose magni-
tude is the same as the characteristic impedance but has a small
reactive component. This suggests adding a compensating reactance
EXPONENTIAL TRANSMISSION LINE 563
to the resistance load. From (3) the best single reactive element is
found to be a condenser whose impedance is equal to the impedance
level at the cutoff frequency. This gives a value oi K — 1 — jv which
when substituted in (5) shows that the input impedance is to a first
approximation a constant times the terminal impedance. To correct
for the reactive component of the input impedance an inductance
having an impedance jZi/v which is equal to the input impedance level
at cutoff is shunted across the input. The resulting impedance trans-
forming network consists of an exponential line with a series capaci-
tance at the high impedance end and a shunt inductance at the low
impedance end. When terminated in a resistance load at either end
equal to the impedance level at that end the input impedance, to a
first approximation, is a resistance equal to the impedance level at the
input end. In fact the deviations of the input impedance from the
ideal for transmission in one direction are just the reciprocal of those
for transmission in the other direction.
The magnitudes of the series capacitance and shunt inductance that
give the improved network may be expressed in terms of the electro-
static capacitance and loop inductance of the line. Simple calculation
shows that the required series capacitance is equal to 2/(^ — 1) times
the electrostatic capacitance of the line and the required shunt in-
ductance is equal to the same factor times the total inductance of
the line.
There is an interesting relationship between these terminations and
a simple high-pass filter. The LC product of the shunt and series arms
of the filter resonates at /i. If an ideal transformer with transforma-
tion ratio k is inserted between the shunt inductance and the series
capacitance, the capacitance becomes Cjk and the new LC resonates
at /iV^. This is the same frequency at which the series capacitance
and shunt inductance that are added to the terminations of the ex-
ponential line resonate. Furthermore the reactance of the shunt
inductance is equal to the impedance level at the cutoff frequency and
the reactance of the series capacitance is equal to the impedance level
at the cutoff frequency exactly as in the case of the high-pass filter.
By using the exponential line it is possible to construct a network
with properties that no network with lumped circuit elements possesses,
namely, a high-pass impedance transforming filter.
Critical Lengths
Besides the characteristics of the exponential line that are sub-
stantially independent of the length of the line, it has properties that
564 BELL SYSTEM TECHNICAL JOURNAL
depend on the length of the line that are analogous to those of a uniform
line a half wave-length or quarter wave-length long. For non-
dissipative lines above the cutoff frequency (5) becomes
i^cos(|-2^)-fisin^
Z, jY T-^ Z. (8)
cos(^^-H2M+jXsin|
When the line is an integral number of half wave-lengths long (27? = tt)
this reduces to
£i = KZ, = kZ2, (9)
which says that the input impedance is equal to the impedance trans-
formation ratio times the load impedance. The length of exponential
line that corresponds to a quarter-wave uniform line differs from an odd
multiple of a quarter wave-length by an amount such that
/ rj - (2« + 1)t\ K^ - I
tan i ^ 2 ) " K'+ 1 ^' ^ ^^
for which (8) becomes
Z,=^^ (11)
Similar expressions exist for the step-down line, but l/iC must be
substituted for K in (10) for the length corresponding to the quarter-
wave uniform line.
With Dissipation ^
An exponential line is an improvement over the uniform "iron wire"
line as a resistance load that will dissipate a large amount of power.
Provided the attenuation is not too large the current and voltage
distribution will be the same as for a non-dissipative line except for
the additional power loss so that we may use the equations for an
exponential line even though the distributed series resistance and
shunt leakage do not vary exponentially with distance.
Suppose that the conductor size and resistance that will just dissi-
pate the desired input power result in an attenuation constant ao for
a uniform transmission line. To a first approximation the conductors
can carry the same current irrespective of the impedance level. The
current wave will be given by the first term of equation (2) which
becomes
EXPONENTIAL TRANSMISSION LINE 565
except for a phase factor. In order that the current will not increase,
5 = — 2a. The actual attenuation "constant,"
will increase with distance down the line so that the current will
decrease but not as rapidly as with a uniform line. The total attenua-
tion in nepers is approximately
1 /'\ r' o„., /i , 1 P\/e'"oi - 1
1 +-,U a^'-'-dx = 1 +-i- 1 ^— — ^ . (14)
At the point where the attenuation of the uniform line is 6 db the
tapered line has an additional attenuation of 7 db above the uniform
Une or a total attenuation of more than twice. The current has been
reduced to less than half. Here an improvement may be made by
increasing the dissipation by either changing the wire size or resistivity
of the conductor. A greater improvement would result from changing
the resistivity because then the capacity for heat dissipation would be
the same. Suppose, however, that one conductor material is to be
used throughout and the dissipation capacity is proportional to the
wire surface; then at this point the wire size could be reduced to 1/2,
doubling the attenuation factor. It is already 4 times that for the
uniform line, so this increases it to 8 times. The resulting total attenua-
tion is 30 db in a length that would have less than 7 db if the line were
uniform. If this attenuation were required the length of line could be
reduced by a factor of about 4.4. Of course the spacing is very close
at the end of this line, but the line could be shorted at the end. This
would approximately double the current at the end, but here again the
current carrying capacity of the line is more than double the current
traveling down the line. With the line shorted the reflected current
would be 60 db down, which would not affect the input impedance
appreciably. For the first 13 db of attenuation the impedance of the
line would be relatively free from changes due to changes in spacing
resulting from wind, etc. When the spacing is small enough to be
affected by wind, vibration, etc., the attenuation will be great enough
to suppress these small irregularities.
Experiment
In order to verify the foregoing theoretical development, measure-
ments have been made on several experimental lines. Figure 4 shows
the results of measurements on two such lines. These lines were
566
BELL SYSTEM TECHNICAL JOURNAL
constructed of No. 12 tinned copper. ' At the^low impedance end the
strain was taken by a victron insulator which also served as a line
spreader and terminal mounting. At the high impedance end the
strain was taken by 1/4" manila rope without other insulation. The
line spacing was adjusted by "lock stitch" tension insulators spaced
1 meter and 1/2 meter apart on the low and high impedance end re-
spectively of the 9-meter line. The 3-meter line was supported at the
1/4, 1/2, 2/3, 3/4 and 7/8th points.
The impedance was measured by the substitution method. To
facilitate the substitution of the reactive component of the line it was
bridged by an antiresonant circuit. Pencil leads calibrated on direct
current] were used as the resistance standards. Type BW IRC 1/2
^
1
>* *
* •
•
i •
y
* — ,
• '
-"^
0^
n° A
*•
■A
0.3
f/fo
0.8
Fig. 4 — Input impedance characteristic. Comparison of theoretical curve with
experimental points for 600 : 300 ohm lines.
Solid circles — 9-meter line.
Open circles — 3 -meter line.
watt resistances were used for terminations. The solid circles of Fig. 4
represent measurements on the 9-meter line. The agreement with
theory is as good as is usually found for actual "uniform lines." In
order to check the theory further toward the lower frequency end —
beyond the range of the measuring equipment — -measurements were
made on a 3-meter line. These measurements are shown by the open
circles. The agreement with theory is not so good, but here the lengths
of the connecting leads are an appreciable fraction of the length of
the line.
Preliminary tests on a full size model of exponential line impedance
transformer showed deviations from the theoretical that might be at-
tributed to improper termination, irregularities along the line, irregu-
larities introduced at the change in conductor size or capacitance of the
spacing insulators. Since it was impossible to determine which of
these was the predominant cause of the deviations from the ideal, it
EXPONENTIAL TRANSMISSION LINE
567
was decided to introduce each of these factors one at a time. This test
was made on a 600 : 300 ohm Hne constructed of No. 6 copper wire with
lockstitch insulators except at the terminals. The correct termination
was obtained by tests on a uniform 300 ohm line with the same physical
structure at the termination. Of necessity the tying of the wire to the
strain insulators at the end introduced a shunt capacitance which
augmented the inherent additional capacitance due the "end effect."
This additional capacitance is equal to that of a short length of line.
Fig. 5 — Photographs of terminations of 300 ohm line,
upper right for curve 1 1
lower for curve 2 ^ of Fig. 6.
upper left for curve 3 J
If the correct amount of inductance is inserted in series with the
resistance load the combined effect of the additional capacitance and
inductance becomes the same as the addition of a small length of line
for all frequencies up to those for which this length of line is an appre-
ciable fraction of a wave-length. Accordingly a small amount of induc-
tance was inserted in series with the resistance as shown in the right
picture of Fig. 5. The input impedance of the uniform line with this
termination is given by "Experimental Curve 1 " at the bottom of Fig. 6.
568
BELL SYSTEM TECHNICAL JOURNAL
A three-inch length of No. 18 wire was inserted as shown in the lower
picture of Fig. 5 and " Experimental Curve 2 " resulted. This reduced
the irregularities in the input impedance to about half, so another three
264
10 12 14 16 18 20 22 24 26 28 30
FREQUENCY IN MEGACYCLES
Fig. 6 — Lower. Experimental input impedance characteristics of 300 ohm line
with terminations shown in Fig. 5. Upper. Input impedance characteristics of
50-meter 600 : 300 ohm line of No. 6 conductors.
inches were inserted, resulting in "Experimental Curve 3." Here the
maxima and minima are displaced, indicating that the effect of the
stray capacitance has been reduced to the same order of magnitude as
that due to the deviation of the resistance from the desired value. This
EXPONENTIAL TRANSMISSION LINE
569
termination was accordingly removed to the exponential line, resulting
in the "Experimental Curve" at the top of Fig. 6. It agrees within
experimental error with the "Theoretical Curve." The slight vertical
displacement of the experimental curve at the higher frequencies is
attributed to deviations in the impedance of the pencil lead, which was
Fig. 7 — Photograph of one of the changes in conductor size.
used as a resistance standard, from a pure resistance equal to its direct
current value.
To increase the power carrying capacity of the exponential line, one
was built with larger wire size at the lower impedance end. This
increased the breakdown voltage by increasing the spacing and con-
ductor diameter and at the same time increased the current carrying
capacity by decreasing the resistance and increasing the heat dissipat-
570
BELL SYSTEM TECHNICAL JOURNAL
ing capacity of the conductors. This was a 600 : 300 ohm Hne con-
structed of 20 meters No. 6 wire, 10 meters 1/4" tubing and 20 meters
3/8" tubing. Here again the correct termination was determined by
measurements on a 300 ohm uniform line of 3/8" tubing. The total
length of terminating loop that gave the best termination was 6)^"
in this case compared with lOj^" for the 300 ohm line of No. 6 wire.
Since no attempt was made to reduce the variations in input impedance
to less than ± 1 per cent these lengths may be as much as an inch off.
These measurements indicated that the exponential line would per-
form satisfactorily as an impedance transformer if it could be con-
structed to have the desired mechanical features without impairing its
electrical properties. The greatest difficulty appeared to reside in the
1.06
FREQUENCY IN MEGACYCLES
P'ig. 8 — Input impedance characteristics of 50-meter 600 : 300 ohm
line of 3/8", 1/4" and No. 6 conductors.
insulators. Special isolantite insulators were designed that would be
satisfactory commercially and still keep the additional capacity to a
reasonable value. Figure 7 shows the construction of the line at the
supporting poles where the conductor size changes.
The results of measurements on this line are shown in Fig. 8. The
solid curve was calculated from the equations developed earlier. The
two broken curves are the results of measurements on the line, one
without insulators and one with insulators. While the insulators affect
the line somewhat they do not increase the deviation from the ideal
appreciably. [The improvement in the agreement between experiment
and theory in this set of curves over that in Fig. 4 is presumably due
to the fact that the comparison resistance for Fig. 8 consisted of 3-IRC
EXPONENTIAL TRANSMISSION LINE
571
resistances instead of the pencil lead. With the fixed IRC resistance it
was, of course, impossible to adjust the standard to exactly the same
value as the unknown. In this case the small difference was determined
by using the slope of the rectifier voltmeter calibration.] This line has
a maximum deviation from the desired input impedance of ± 6 percent
for all frequencies above 4.2 mc. (Measurements were made up to 28
mc.) The phase angle of the input impedance was found to be zero
10 12 14 16 18 20 22 24 26 28
FREQUENCY IN MEGACYCLES
Fig. 9 — Input impedance characteristics of 15-meter 600 : 300 ohm
line of No. 6 conductors
within the accuracy of measurement. From theory the phase angle
would be expected to vary between — 0° and + 3°.
The curves of Fig. 9 refer to a 600 : 300 ohm line of No. 6 wire 15
meters long. With resistance termination this line has rather large
variations in the input impedance but with the addition of the proper
reactances the input impedance is flatter than the longer line with
resistance termination. At the lower frequencies where the variations
in the input impedance were large without the reactive networks, their
addition gives approximately the expected improvement. At the
higher frequencies the inductance was approximately anti-resonated
572 BELL SYSTEM TECHNICAL JOURNAL
by its distributed capacity and the input impedance approaches that
for the resistance termination.
Conclusion
Theory indicates that the exponential Hne may be used as an imped-
ance transformer over a wide frequency range. The results of experi-
ment show that the desired characteristic'can be realized in practice.
Among the applications of the exponential line may be mentioned its
use in transforming the impedance level back to its original value after
the paralleling of two transmission lines feeding two antennas. It
could be used to transform the input impedance of a rhombic antenna
down to the usual 600-ohm level of open wire transmission lines. If
twin coaxial lines are used inside the transmitter building to eliminate
undesired feedback, coupling, etc., the exponential line could be used to
transform from the highest practical impedance level of such lines to a
practical level of the more economical open wire lines for use outside
the building.
Appendix
The exponential line is a non-uniform line so that the terms "charac-
teristic impedance" and "surge impedance" of an exponential line are
not synonymous. The terms "surge impedance" ' and "nominal
characteristic impedance" ^ may be used synonymously for the charac-
teristic impedance of the uniform line that has the same distributed
constants as the non-uniform line at the point in question. Expressed
'as functions of the distributed "constants" of the line they are the
square root of the ratio of the distributed series impedance to the
distributed shunt admittance at the point along the line in question.
It will be expedient to refer to the nominal characteristic impedance
as the impedance level at the point in question. Schelkunoff ^ has
defined the characteristic impedances as the ratio of voltage to current
at the point in question for each of the two traveling waves of which
iThe term "surge impedance" is defined by A. E. Kennelly on page 73 of "The
Applications of Hyperbolic Functions to Electrical Engineering Problems " (McGraw-
Hill 1916) as follows: "The surge impedance of the line is not only the natural imped-
ance which it offers everywhere to surges of the frequency considered, but it is also the
initial impedance of the line at the sending end." Hence the "surge impedance"
should be independent of the configuration of the line except at the point in question
and in particular it should be equal to that for a uniform line constructed so as to have
the same dimensions everywhere as the non-uniform line has at the point in question.
2 The word nominal as used here has the same meaning as in "nominal iterative
impedance" as used by K. S. Johnson in "transmission circuits for telephone com-
munication" (Van Nostrand 1925). ,
3 S. A. Schelkunoff, "The Impedance Concept and its Application to Problems of
Reflection, Refraction, Shielding and Power Absorption," Bell System Technical
Journal, 17, 17-48, January, 1938.
EXPONENTIAL TRANSMISSION LINE 573
the steady state condition is composed. At each point an exponential
Hne has two characteristic impedances which are different and depend
upon the frequency as well as the position along the line.
Because of the change of impedance level, the propagation constants
for the voltage and current differ, so that it is convenient to consider
the transfer constant ^ which may be defined as half the sum of the
voltage and current propagation constants.
* Compare with the definition of "image transfer constant" as given by K. S.
Johnson in "Transmission Circuits for Telephone Communication."
The Bridge Stabilized Oscillator*
By L. A. MEACHAM
A new type of constant frequency oscillator of very high stability is
presented. The frequency controlling resonant element is used as one arm
of a Wheatstone resistance bridge. Kept in balance automatically by a
thermally controlled arm, this bridge provides constancy of output ampli-
tude, purity of wave form, and stabilization against fluctuations in power
supply or changes in circuit elements. A simple one-tube circuit has
operated consistently with no short-time frequency variations greater than
± 2 parts in 10^. Convenient means are provided for making precision
adjustments over a narrow range of frequencies to compensate for long-time
aging effects.
Description of the circuit is followed by a brief linear analysis and an
account of experimental results. Operating records are given for a 100 kc.
oscillator.
Introduction
THE problem of improving the stability of constant frequency
oscillators may be divided conveniently into two parts, one
relating to the frequency controlling resonant element or circuit, and
the other to the means for supplying energy to sustain oscillations.
The ideal control element would be a high-(3 electrical resonant circuit,
or a mechanical resonator such as a tuning fork or crystal, whose
properties were exactly constant, unaffected by atmospheric conditions,
jar, amplitude of oscillation, age, or any other possible parameter.
The ideal driving circuit would take full advantage of the resonator's
constancy by causing it to oscillate at a stable amplitude and at a
frequency determined completely by the resonator itself, regardless of
power supply variations, aging of vacuum tubes or other circuit ele-
ments, or the changing of any other operating condition.
This paper, concerning itself principally with the second part of the
problem, describes an oscillator circuit which attains a very close
approximation to the latter objective. The "Bridge Stabilized Oscil-
lator" provides both frequency and amplitude stabilization, and as it
operates with no tube overloading, it has the added merit of delivering
a very pure sinusoidal output.
Oscillator Circuit
The bridge stabilized oscillator circuit, shown schematically in Fig. 1,
consists of an amplifier and a Wheatstone bridge. The amplifier out-
* Presented at Thirteenth Annual Convention of Institute of Radio Engineers,
New York City, June 16, 1938. Published in Proc. I. R. E., October 1938.
574
THE BRIDGE STABILIZED OSCILLATOR
575
put is impressed across one of the diagonals of the bridge, and the
unbalance potential, appearing across the conjugate diagonal, is applied
to the amplifier input terminals. One of the four bridge arms, Ri, is a
thermally controlled resistance; two others, Ri and Rz, are fixed re-
sistances, and the fourth, Zi = Ri -\- jXi, is the frequency-controlling
resonant element.
In this discussion Zi is assumed to represent a crystal suitable for
operation at its low-impedance or series resonance. A coil and con-
denser in series could be substituted, and even a parallel-resonant
control element might be used by exchanging its position in the bridge
VOLTAGE ATTENUATION
VOLTAGE AMPLIFICATION
Fig. 1 — Schematic circuit diagram of bridge stabilized oscillator.
with i?2 or R3. Operating a crystal at series resonance has the advan-
tage of minimizing effects of stray capacitance.
The bridge is kept as nearly in exact balance as possible. Assuming
that Ri, Ri and Rz are pure resistances, we may write for exact reactive
balance,
Xi = 0,
and for exact resistive balance,
-^1 _ -^3
R2 Ri
In order that the circuit may oscillate, a slight unbalance is required.
Accordingly i?i must be given a value slightly smaller than {RzRzj/Ri,
576 BELL SYSTEM TECHNICAL JOURNAL
so that the attenuation through the bridge is just equal to the gain of
the ampUfier.
It is evident that if all the bridge arms had fixed values of resistance,
the attenuation of the bridge would be very critical with slight changes
in any arm. This would obviously be undesirable, for the circuit
would either fail to oscillate, or else build up in amplitude until tube
overloading occurred. The thermally controlled resistance Ri elimi-
nates this difficulty. This arm has a large positive temperature coeffi-
cient of resistance, and is so designed that the portion of the amplifi.er
output which reaches it in the bridge circuit is great enough to raise its
temperature and increase its resistance materially. A small tungsten-
filament lamp of low wattage rating has been found suitable. It
functions as follows:
When battery is first applied to the amplifier, the lamp Ri is cold and
its resistance is considerably smaller than the balance value. Thus
the attenuation of the bridge is relatively small, and oscillation builds
up rapidly. As the lamp filament warms, its resistance approaches
the value for which the loss through the bridge equals the gain of the
amplifier. If for some reason Ri acquires too large a resistance, the
unbalance potential e becomes too small or possibly even inverted in
phase, so that the amplitude decreases until the proper equilibrium is
reached.
This automatic adjustment stabilizes the amplitude, for the amount
of power needed to give Ri a value closely approaching {RiRzjjRi is
always very nearly the same. A change in the amplifier gain would
cause a readjustment of the bridge balance, but the resulting variation
in R\ or in the amplifier output would be extremely small. The
operating temperature of the lamp filament is made high enough so
that variations in the ambient temperature do not affect the adjust-
ment appreciably.
No overloading occurs in the amplifier, which operates on a strictly
Class A basis, nor is any non-linearity necessary in the system other
than the thermal non-linearity of Ri. As the lamp resistance does not
vary appreciably during a high-frequency cycle, it is not a source of
harmonics (or of their intermodulation, which Llewellyn ^ has shown
to be one of the factors contributing to frequency instability).
In contrast to the lamp, an ordinary non-linear resistance, of copper
oxide for example, would not be suitable for Ri. A resistance of the
thermally-controlled type having a negative temperature coefficient
1" Constant Frequency Oscillators," F. B. Llewellyn, Proc. I. R. E., December
1931.
THE BRIDGE STABILIZED OSCILLATOR 577
could be used by merely exchanging its position in the bridge with
i?2 or Rz.
The frequency control exerted by the crystal depends upon the fact
that the phase shift of the amplifier must be equal and opposite to that
through the bridge. In the notation of Black,^ applied to the circuit
of Fig. 1,
E , ,,„
^=7
= IMI
The condition for oscillation is
M/^ = 1 l_0,
which implies that | /i|3 1 = 1 and d = — \]/.
The vector diagrams of Fig. 2 illustrate the frequency-stabilizing
action of the bridge by showing the voltage relations therein for two
values of amplifier phase shift, 6. When d is zero, as in diagram A,
the unbalance vector e is in phase with the generated voltage E applied
to the bridge input, and thus all the vectors shown are parallel. They
are displaced vertically from each other merely to clarify the drawing.
The crystal is here constrained to operate at exact resonance.
In diagram B, the amplifier is assumed to have changed its phase for
some reason by an amount far in excess of what would be anticipated
in practice, 6 hene having a value of + 45 degrees. The important point
to be observed is that the ratio of d to the resulting change in the phase
angle <^ of the crystal impedance Z4 is very large. That is, the crystal
is still operating close to resonance in spite of the exaggerated change in
the driving circuit. If the gain of the amplifier were greater, the action
of the thermally controlled resistance would keep the amplifier output
vector E practically the same in length, making the unbalance vector e
correspondingly shorter. The angle 4> would therefore have to be more
acute for the same value of 6, and it follows that with increased gain the
crystal is held closer to true resonance and the stability is improved.
When 6 equals zero, changes in | /u | do not affect the crystal operating
phase, but for any other small value of d, gain variations cause slight
readjustments of the angles between vectors. The amplifier should
accordingly be designed for zero phase shift, and also, of course, should
have as much phase stability as possible.
^ "Stabilized Feedback Amplifiers," H. S. Black, Bell System Technical Journal,
January 1934.
578
BELL SYSTEM TECHNICAL JOURNAL
In this discussion the input and output impedances of the amplifier,
i?5 and Rs, are assumed to be constant pure resistances. Actually,
changes in the tube parameters or in certain circuit elements are likely
to affect both the magnitude and the phase of these impedances. It
may be shown, however, that such changes have the same effect upon
the bridge and upon the frequency as do changes of about the same
I4Z4 ^
I2R2
I3R3 ,
I1R1 ^
|J-1.^5_1_
e = o
\
uz.
\
A- -f^ ^ E' 11
LOCUS OF TAIL OF
VECTOR e FOR
VARYING FREQUENCY
•F — O
Fig. 2 — ^Vector diagrams illustrating operation of bridge oscillator, with simplify-
ing assumptions that R5 is large and that E and E' are strictly in phase.
A — At resonance
Z4 = Ri+jO
d = 0
i?l < i?2 = i?3 = R4
B — Above resonance
Z4 = i?4 + jXi
X4 Inductive
0 = + 45°
i?l < i?2 = i?3 = -R4« R&
percentage in [mI or 0; therefore all variations in the driving circuit
external to the bridge may be assumed for convenience to be repre-
sented by variations in its gain and phase.
This leniency with regard to R5 and Re does not apply to the other
bridge resistances, however. Ri, R2 and R3 are directly responsible
for the crystal's operating phase and amplitude; they should be made
as stable and as free from stray reactance as possible.
THE BRIDGE STABILIZED OSCILLATOR
579
The effect of the bridge upon harmonics of the oscillation frequency
is of interest. Harmonics, being far from the resonant frequency of the
crystal, are passed through the bridge with little attenuation but with a
phase reversal approximating 180 degrees, as illustrated by the dotted
locus in Fig. 2. Thus if the amplifier were designed to cover a band
broad enough to include one or more harmonics and if care were taken
to avoid singing at some unwanted frequency, a considerable amount
of negative feedback could be applied to the suppression of the har-
monics in question.
Circuit Analysis
In the following section, expressions are derived for the frequency
of oscillation in terms of the gain and phase shift of the amplifier, the
Q of the crystal, and values of the bridge resistances. It is assumed
that the latter are constant and non-reactive, and therefore, as ex-
plained previously, that all sources of frequency fluctuations apart from
changes in the crystal itself appear as variations in | ;ti | or ^. Because
the bridge oscillator does not rely upon non-linearity in the ordinary
sense to limit its amplitude, the analysis can be based reasonably on
simple linear theory.
In the near vicinity of series resonance the crystal may be repre-
sented accurately by a resistance Ra, inductance L and capacitance C,
connected in series. The reactive component of the crystal's im-
pedance is accordingly
Solving for the frequency,
'LC - 1
(1)
2L^
1
IL ^ LC
\X,
LC[ 2
C
^Lc[
1 +
L
X,
+ \1 +
m-
C 1/X_4
L~^2\ 2
C
L
C\2
_1 1/X_4 jC
2' 4\ 2 \L
+
(2)
Near series resonance, (X4/2) ^{CIL) < < 1. We therefore disregard
powers higher than the first in the series expansion above and obtain
the close approximation.
580
BELL SYSTEM TECHNICAL JOURNAL
The frequency deviation from resonance, expressed as a fraction of
the resonant frequency /o, is thus
/ — /o CO — COo , A'4 \C
/o
Wo
(4)
and in the region of interest, where coL and 1/wC are approximately
equal,
f — fo . Xi -^4
/o
2coL 2QRi
(5)
Considering now the bridge circuit, and applying well-known equa-
tions,^ we obtain
hRr, _ AR, - jBX,
E ~ MRi + jNXi '
(6)
^hich
and
A - R.iR^Rz - RiRa),
B = RiRiRc,,
M = {R, -f R2)(R,Ra + RM + (Rs + Ra){RiR2 + RM
+ (i?5 + R6){RlRi + i?2i?3) + i?5(i?1^3 + R2Ri)
+ 7?6(i?li^2 + RsRi),
N = R,{Ri + R, + i?5)(i?2 + i^e) + RiRi{Rs + i^s).
(7)
The condition for oscillation, as mentioned previously, is jjl^ = 1|0.
Putting /x = Ml + iM2, we may write
(mi + JM2)
ARi - jBXi
AIRi + JWX4
which gives the pair of equations
fiiARi + ^2^X4 - MRi = 0
HiARi - {fiiB + N)Xi = 0.
1,
and
(8)
(9)
(10)
For the special case in which the amplifier phase shift is zero (m = 0),
these become
Ml = ^ = IMI
(ii:
and
X4 = 0. (12)
3 "Transmission Circuits for Telephonic Communication," K. S. Johnson, pp.
284-5. D. Van Nostrand Company.
THE BRIDGE STABILIZED OSCILLATOR
581
The latter equation indicates that the frequency is then independent
of changes in any of the circuit parameters except the crystal, which
must operate exactly at resonance.
If the phase of ix differs only slightly from zero, so that jU2 is very
small, then it may be inferred from continuity considerations that the
frequency is still very nearly independent of all circuit parameters,
except of course variations in d, the phase of y.. When Q is limited to
values for which ^2^X4 < < iiiARi, (11) still applies closely. Substi-
tution into (10) gives
X4 =
MR A
B,xr + N
MRS
B
M
+ iV
(13)
and finally from (5) and (13) we obtain the frequency deviation in the
form
/ - /o . Md
/o
2Q{B\tx\ +N)
(14)
As noted above, this expression applies accurately only when 6 is
small, as it should be in a well designed bridge oscillator.
The effect of variations in the amplifier may be examined by dif-
ferentiating (14). For changes in d only,
and for those of | /x | ,
4/1
M
/oj
, 1
e~ 2Q{B\y.\ +iV)'^
IX|,
dn
iMi
BMd
/oJ
2(2(5 ImI +Ny
dd,
d\
(15)
(16)
Equations (15) and (16) have been found to be closely in accord with
experiment, although the differentiation is not rigorously allowable
(B, M and N being only approximately constant).
In the special case where all the fixed bridge resistances (R2 to Re
Inclusive) are equal, and |m| is large enough so that Ri has nearly the
same value, (14), (15) and (16) reduce to the following:
/-/o
/o
df]
/oJ
df]
/oJ
iMl
<2(ImI +8)
8
e <2(|m|+8)
dd,
<2(|mI +8)^
dU\
(17)
(18)
(19)
582 BELL SYSTEM TECHNICAL JOURNAL
These expressions show, as did the vector diagrams, that for optimum
stability the amplifier phase shift should be made approximately zero,
the crystal should have as large a value of Q (as low a decrement) as
possible, and the amplifier should have high gain. Linearity in the
amplifier is also desirable, to minimize the modulation effects described
by Llewellyn.' When present, these effects appear as variations in
ImI and d.
One of the significant differences between the bridge oscillator and
other oscillator circuits is the fact that its frequency stability is roughly
proportional to \ix\. This relationship holds at least for amounts of
gain that can be dealt with conveniently. Although increased gain is
generally accompanied by larger variations in phase, the two are not
necessarily proportional. For example, if greater stability were re-
quired for some precision application than could be achieved with a
single-tube bridge oscillator, and if the constancy of the crystal itself
warranted further circuit stabilization, it could be obtained by adding
another stage. The phase fluctuations in the amplifier might possibly
be doubled, but the value of ] ju I would be multiplied by the amplifica-
tion of the added tube, giving an overall improvement.
To illustrate the high order of stability provided by a bridge oscil-
lator, let us consider a model composed of a single-tube amplifier and
a bridge in which all the fixed resistances are made equal to that of the
crystal. We will assume the crystal to have a reasonably high ^ Q
of 100,000. The amplifier phase, let us say, is normally zero, but may
possibly vary ± 0.1 radian (±6 degrees), and the value of |/x|,
nominally 400, maychange ±10 per cent. From (18) and (19) we find
A/
/o
_ (8)(0-l) - ± 2 17 X 10-
r ^ (100,000) (360 + 8) " ^ ^-'^ ^ '^
and (when Q has its maximum value of 0.1 radian)
A/
/o
(8) (0.1) (40) _ ^ 2 36 X 10-«
^ (100,000) (360 -f 8)2 " ^ ^-^^ ^ ^^ •
This example represents the degree of stabilization against circuit
fluctuations that can be obtained with a simple form of the oscillator
operating under poorly controlled conditions. By stabilizing the power
supply and other factors affecting | ^ | and d, and by increasing the gain,
the frequency variations arising in the driving circuit may be made
negligible compared to the variations found at present in the properties
even of the best mounted crystals.
* For crystals in vacuum, values of Q as great as 300,000 have been obtained.
THE BRIDGE STABILIZED OSCILLATOR
583
Experiment
The circuit diagram of an experimental bridge stabilized oscillator is
shown in Fig. 3, and its photograph in Fig. 4. The amplifier unit
consists of a single high-mu tube Fi with tuned input and output
transformers T\ and Ti and the usual power supply and biasing
arrangements. The crystal, mounted in the cylindrical container at
the left end of the panel, is one having a very low temperature coeffi-
cient of frequency at ordinary ambient temperatures. In Fig. 4 it is
shown without provisions for temperature control. A high Q is
obtained by clamping the crystal firmly at the center of its aluminum-
TO
8 < LOAD
Fig. 3 — Circuit of experimental bridge oscillator.
coated major faces between small metal electrodes ground to fit, and by
evacuating the container.
Some of the circuit parameters are listed below :
Ri = Tungsten-filament lamp,
Ri = 100 ohms,
i?3 = 150 ohms,
Zi = 100 kc. crystal.
Characteristics at resonance :
Ri = 114 ohms,
Xl = Xc = 11,900,000 ohms,
Q = 104,000,
R5 = Re — 150 ohms (approx.),
Ri = 500 ohms,
Rs = 200 ohms,
\n\ = 422 (52.5 db voltage gain from e to E).
584
BELL SYSTEM TECHNICAL JOURNAL
Fig. 4-
-Experimental bridge stabilized oscillator without provision
for temperature control.
Figure 5 shows the resistance of the lamp Ri plotted against the
power dissipated in its filament. The large rise in resistance for small
amounts of power is due to the efifective thermal insulation provided
by the vacuum surrounding the filament and to low heat loss by radia-
tion. The lamp operates at temperatures below its glow point, assur-
ing an extremely long life for the filament.
23 4-56 789
POWER INTO LAMP IN MILLIWATTS
Fig. 5 — Characteristic of lamp used for Ri.
THE BRIDGE STABILIZED OSCILLATOR
585
The particular value assumed by Ri in the circuit of Fig. 3 is ap-
proximately (R2Rz)/Ri = [(100)(150)]/114 = 131.6 ohms, and hence
from Fig. 5 it follows that the power supplied to the lamp is about 3.7
milliwatts. The r.m.s. voltage across the lamp is computed to be
0.70 volt, and across the entire bridge, 1.23 volt. The power supplied
to a load of 150 ohms through the pad composed of R-; and Rs is accord-
ingly 0.22 milliwatt, or 6.6 db below 1 milliwatt, which is in agreement
with measurements shown in Figs. 8 and 9, described below. These
quantities are given to illustrate the fact that currents and voltages in
K^b
-—
»
a ,
1
NORMAL
t- OPERATING
POINT
'' i
1
n
1
»
-• — <
c,^
60 80 100 120 140 160 180 200 220 240 260
PLATE BATTERY POTENTIAL IN VOLTS
Fig. 6 — Oscillator frequency vs. plate battery potential.
a — Ci and C2 tuned for maximum amplifier gain.
b — Ci and C2 decreased 5%.
c — Ci and Co increased 5%.
this type of oscillator may be calculated readily from the values of the
circuit elements, and without reference to the power supply voltages or
the tube characteristics except to assume that they give the amplifier
sufficient gain to operate the bridge near balance, and that tube over-
loading does not occur at the operating level.
Experimental performance curves for the circuit of Fig. 3 are pre-
sented in Figs. 6 to 11 inclusive. Figure 6 shows frequency deviation
plotted against plate battery voltage for several settings of the grid
and plate tuning condensers. For curve a the amplifier was adjusted
at maximum gain, corresponding approximately to zero phase shift as
586
BELL SYSTEM TECHNICAL JOURNAL
well. Here the frequency varied not more than one part in one hun-
dred million for a voltage range from 120 to 240 volts. Curve h was
taken with the two tuning capacitances C\ and Ci decreased 5 per cent
from their optimum settings, and curve c with both capacitances
increased 5 per cent. These detunings introduced phase shifts of
about ± 40° (± 0.70 radian), decreased \)x\ by 0.8 dh and changed
the frequency, as shown in Fig. 6, approximately ± 2 parts in ten
million. Although the analysis should not be expected to apply
0.1
O - 0.2
b
r 1
(
1
, a ,
(
, ^— .
1
-• — NORMAL
OPERATING
POINT
7 8. 9 10 11
FILAMENT BATTERY POTENTIAL IN VOLTS
Fig. 7 — Oscillator frequency vs. filament battery potential.
a — Ci and d tuned for maximum amplifier gain.
b — Ci and d decreased 5 %.
c — Ci and Ci increased 5%.
accurately for such large phase shifts, calculation of the frequency
deviations by means of (18) gives ±1.4 parts in ten million — in fair
agreement with the experimental results. As might be expected,
curves h and c show somewhat less stability with battery voltage
changes than does curve a.
Figure 7 presents a similar set of curves for variations of filament
voltage. Here, for the "maximum-gain" tuning adjustment, a drop
from 10 volts, the normal value, to 8 volts caused less than one part in
one hundred million change of frequency, as shown in curve a.
THE BRIDGE STABILIZED OSCILLATOR
587
In Fig. 8, the gain of the amplifier and the output level of the oscil-
lator are plotted against plate battery voltage, while in Fig. 9 the same
quantities are related to filament potential. These curves show that
although power supply variations change the amplifier gain, they have
but slight effect upon the amplitude of oscillation. This stabilization
is produced, as explained heretofore, by the action of the lamp.
The oscillator was designed to work into a load of 150 ohms, its
output impedance approximately matching this value. It might be
expected that variations in the magnitude or phase angle of the load
(D 53
O
> 51
OSCILLATOR OUTPUT LEVEL
/ INTO 150-OHM LOAD
1
,^
_^.
1
H— 1 '
1
1
1
»
1
NOR
OPER>
PC
VIAL
1
,1
^
NT
1 >
^
1
^/^
<
1
A
A
^VOLTAGE AMPLIFICATION OF
VACUUM TUBE CIRCUIT
= 20 LOG,o|^x|
/
1
r
*
■7^,
-8 S
60
80
100 120 140 160 180 200 220
PLATE BATTERY POTENTIAL IN VOLTS
240 260
-10
Fig. 8 — Amplifier gain and oscillator output level vs. plate battery potential.
would affect the frequency materially even though a certain amount of
isolation is provided by i?7 and R^. However, measurements made
with (1) a series of load impedances having a constant absolute magni-
tude of 150 ohms but with phase angles varying between — 90° and
+ 90° and (2) a series of resistive loads varying between 30 ohms and
open circuit, showed less than one part in a hundred million frequency
variation. Graphs of these results have not been included, since they
practically coincide with the axis of zero frequency deviation.
The tuned transformers Ti and T^ in this experimental model pre-
cluded the suppression of harmonics by negative feedback, |/x| being
small at the harmonic frequencies. The tuning itself provided sup-
pression, however, so that the measured levels of the second and third
588
BELL SYSTEM TECHNICAL JOURNAL
harmonics in the output current were respectively 67 db and 80 db
below that of the fundamental. This purity of wave form is of course
largely dependent upon the absence of overloading.
To correct any small initial frequency error of the crystal and to
allow for subsequent aging, a small reactance connected directly in
series with the crystal provides a convenient means of adjusting the
frequency as precisely as it is known. This added reactance may be
considered as modifying either of the reactances in the equivalent
series resonant circuit of the crystal. Figure 10 shows that for a small
53
> 51
OSCILLATOR OUTPUT LEVEL
, INTO 150-OHM LOAD
^
jL.,
\
r-^^'
y
\
VOLTAGE AMPLIFICATION OF
VACUUM TUBE CIRCUIT
= 20LOG,o ||JL| ^
/
A
\
■•— NORMAL
OPERATING
POINT
8 9 10
FILAMENT BATTERY POTENTIAL IN VOLTS
■l^.
-6
\
\
s.
-10
\
\,
\
\,
-20
s
\,
\
\
■600
-400 -200 0 200 400
REACTANCE IN SERIES WITH CRYSTAL IN OHMS
600
Fig. 10 — Frequency of oscillator vs. adjusting reactance.
were obtained over a period of several months. Figure 11 is a photo-
graph of a section of this record. It shows the short-time variations of
both oscillators plus a small amount of scattering caused by the meas-
uring equipment itself. The crystals were temperature-controlled in
separate ovens, and the power was supplied from separate sets of
laboratory batteries controlled to about ± 2 per cent in voltage.
Shielding was ample to avoid any tendency to lock in step.
In addition to these small short-time variations, the oscillators
exhibited a very slow upward drift in frequency, attributed to aging
590
BELL SYSTEM TECHNICAL JOURNAL
of the mounted crystals. This aging decreased in a regular manner
with time, the mean drift of one of the crystals being less than one part
in ten million per month after three months of continuous operation,
«
•
•
6
J'
m
m
m
m
m
•m
m
*
•
4
,3
•
•
«
•
2
1 PM
-
m
m
®
#
12
11
«
•
10
9
8
6
0 10 20 30 40 50 60 70 80 90 100
PARTS IN 108
Fig. 11 — Record of frequency comparison between two bridge stabilized oscil-
lators. Full scale one part in a million. Variations less than ± 2 parts in one
hundred million.
and about a third of this amount after seven months. In most appli-
cations, gradual frequency drift is not objectionable even though the
required accuracy is very high, for readjustment is merely a matter of
setting a calibrated dial.
THE BRIDGE STABILIZED OSCILLATOR 591
Application
The bridge stabilized oscillator promises to become a useful tool in
many commercial fields as well as in certain purely scientific problems
such as time determination and physical and astronomical measure-
ment. It may be used either to increase the frequency precision in
applications where operating conditions are accurately controlled, or
else to make such control unnecessary, affording high stability in spite
of unfavorable conditions.
An interesting application in the field of geophysics has already been
made in the form of a "Crystal Chronometer." This chronometer
consists of a single-tube bridge oscillator, a frequency dividing circuit,
and a synchronous timing motor. It was recently loaned by Bell
Telephone Laboratories to the American Geophysical Union and was
used with the Meinesz gravity-measuring equipment on a submarine
gravity-survey expedition in the West Indies. Although operating
under somewhat adverse conditions of power supply, temperature, and
vibration, it was reported ^ to be more stable than any timing device
previously available, errors in the gravity measurements introduced
by the chronometer being negligibly small.
® "Gravity Measurement on the U. S. S. Barracuda," M. Ewing, and "Crystal
Chronometer Time in Gravity Surveys," A. J. Hoskinson; pp. 66 and 77 rasp..
Transactions of the American Geophysical Union, Part I, 1937.
Effect of Space Charge and Transit Time on the Shot Noise
in Diodes
By A. J. RACK
The theoretical analysis of the effect of space charge upon the
"shot noise" in a planar diode shows that for practically all
operating conditions, the tube noise is equivalent to the thermal
resistance noise of the plate resistance at 0.644 times the cathode
temperature. Noise in diodes of other than planar shapes is
discussed and it is concluded that the same relation holds. It is
shown that transit time produces the same high frequency modi-
fication for both the thermal and shot tube noise, and that the tube
noise is decreased by transit time.
IN the study of noise in vacuum tubes, the effect of the space charge
upon the shot noise has been a subject of considerable interest and
practical importance. Several papers have been written in which it is
shown that the shot noise is decreased by the space charge, and that the
tube noise in a diode with space charge is equivalent to the thermal
resistance noise of the plate impedance at a temperature slightly
greater than half of that of the cathode.^' ^^ ^' * The most compre-
hensive analysis was made by Schottky and Spenke. These authors,
employing a different method from the one here presented, have
obtained the same general conclusions given in this paper, although
they prefer to express the result in the form of a modified shot-noise
equation, whereas for reasons developed below, the writer prefers the
thermal form. The theoretical analysis and discussion presented here
was undertaken to show in more detail the extent of the range of the
operating condition for which the thermal resistance equivalent of tube
noise is valid and to study the effect of transit time upon both the shot
and thermal tube noise.
For convenience, the paper is divided into three parts. In the first
section is given an exact mathematical treatment of the tube noise at
low frequencies in a parallel plane diode for any degree of space charge.
A discussion of the final tube noise equation obtained through this
analysis, and the extension of these results for the planar diode to any
other shape diode is given in Part II, where the presentation is such
that the section may be read independently of the theoretical analysis
in Part I. Through several approximations. Part III treats the effect
of transit time upon tube noise in the planar diode.
592
SHOT NOISE IN DIODES 593
Part I — ^General Low Frequency Analysis
In the development of the general equations for the direct current
in vacuum tubes with space charge, account has been taken of the
fact that the electrons are emitted from the cathode with Maxwellian
velocity distribution. This fact has been verified experimentally by
Germer,^ and the resulting equations for the relation between current
and voltage have been derived and investigated by Fry,^ Langmuir/
and others. In the extension of this analysis to tube noise, it is only
necessary to assume that the number of electrons emitted with any
velocity does not remain constant, but fluctuates with time according
to the well-known laws of probability. In the analysis on this basis,
the frequencies involved will be considered to be sufficiently low so
that any transit time effect is negligible.
Below is given a list of the definitions of various symbols to be used in
the tube noise study of a parallel plane diode. The practical system of
units is employed throughout.
n(Uc)dUc = instantaneous rate of emission per unit area of the cathode
of electrons with initial velocities between Uc and
Uc + duc in the rjc-direction, regardless of the velocity
components in the other directions,
= no(uc)dUc + 8{Uc)dUc,
no(Uc)dUc = average rate of emission of electrons with A:-directed
velocities between Uc and Uc + dUc,
b{iic)duc = instantaneous deviation from average rate of emission,
/ = instantaneous anode current per unit area,
V — instantaneous potential with respect to cathode of a plane
at a distance x from the cathode,
V — instantaneous potential with respect to cathode of the
potential minimum,
u — instantaneous velocity at ric-plane of electrons which had
an initial x-directed velocity of Uc at the cathode,
x' = instantaneous position of potential minimum,
e = charge on electron = — 1.59 X 10~"^^ coulombs,
m = mass of electron — 9.01 X 10~^^ grams,
h = ratio of dyne cms. to joules = 10~^,
e = permittivity of a vacuum in practical units = 8.85 X 10~^*
farads/cm.,
k = Boltzmann's gas constant = 1.372 X 10~^* watts/degree
Kelvin,
A^ — average total number of electrons emitted per second per
unit area from the cathode,
T — absolute temperature of the cathode.
594
BELL SYSTEM TECHNICAL JOURNAL
In the following analysis, it is assumed that the electrodes of the
planar diode are infinite in extent, and that the electron emission is
random, so that the equipotential surfaces are parallel planes perpen-
dicular to the A:-axis.
The potential distribution in such a planar diode operating with
space charge is shown in Fig. 1. The origin of coordinates is taken at
the cathode, and the potential minimum formed by space charge occurs
at a distance x = x' from the cathode. The subscript a will be used to
denote the space between cathode and potential minimum while ^
applies similarly to the space between minimum and anode. Of all
Fig. 1- — Potential distribution in planar diode.
the electrons emitted from the cathode only those whose x-velocity
exceeds the value uj corresponding to the potential minimum can
penetrate the barrier and proceed to the anode. Electrons with lesser
values of initial velocity will come to rest at a point in the a-region
where the potential corresponds to their initial velocity and will then
return to the cathode. The anode current density is thus given by
■f.
I = e i n(u,)duc,
(1)
while the relation between velocity u and potential V at a given value
of X is
2e
u^ = u? —
hm
V.
(2)
SHOT NOISE IN DIODES 595
A third fundamental relation is Poisson's equation which becomes in
the parallel plane case under consideration
In the a-region the total charge density is made up of three classes of
electrons, namely
1. Those destined to pass the potential minimum and arrive at the
anode.
2. Those moving away from the cathode but which will not travel as
far as the minimum point.
3. Those returning to the cathode.
Corresponding to each class of electrons, there is an associated
current, pu, so that each of the three densities pi, P2 or p3, may be
expressed by a relation of the form,
- = ^- w
When it is remembered that the potential and velocity at a given
value of X are uniquely related through (2), then it is easy to see that
the total density for a given plane in the a-region is given by
/ fi(u ) I ' fi(u )
Pa = e \ ■ ~ duc + 2e I ' duc, (5)
where the first term represents the contribution of electrons in class 1
above, while the second term represents the contribution of electrons in
classes 2 and 3. The contribution of class 3 is equal to that of class 2.
The lower integration limit v of the second term of (5) represents the
initial velocity of an electron which would just arrive at the value of x
under consideration before coming to rest and starting back toward the
cathode and the limit u/ in both terms represents the initial velocity of
an electron which comes to rest just at the potential minimum.
Thus, from (2)
J^ V and u/ = J~ r.
(6)
In the j8-region there is only one class of electrons, so that the
density is more simply expressed. Thus,
■
Pe = e ] -^:^duc. . (7)
The value of p in (5) and (7) may each be expressed in terms of
d^V/dx^ by the use of (3), and the integration of these two Poisson's
relations for the common boundary "condition that the electric force is
596 BELL SYSTEM TECHNICAL JOURNAL
zero at the potential minimum has the following result:
, ■ ° — I (m — u')n{uc)duc -\ • I un{uc)dur, (8)
{dx) e Jj,^, e Jp
{dx) e
I (m — u')n{uc)duc, (9)
where u' is the electronic velocity at the potential minimum, i.e.,
{u'Y = Ue- - {2e/hm) V .
At this point the analysis departs for the first time from the classic
analyses of Fry ^ and Langmuir,^ through the introduction of the
concept that the instantaneous rate of emission may be expressed as the
sum of an average rate of emission plus an instantaneous deviation.
That is,
«(Mc) = Wo(m<,-) + 5(Mc), (10)
transforms (8) and (9) into the following equations:
— I (m — ic )n(i{u,)dUc
f ''uc'
{dx)
where
and
where
4hm C""'
H uno{u,)duo + a{d), (11)
e Jo
"Zhttt I AhtH I
a(8) = I (w — u')8{uc)dUc H I u8{Uc)dUc
{dV^ ^2hm r ^^_ ^/)„^(^^)^^^ _^ ^(5)^ (12)
{dx) e J„/
^(5) ^ ^ r (^ _ u')5{Ur)dUr.
e Juc'
Since the average rate of emission may be expressed by the Max-
wellian relation,
«o(«.) = laNUce-""'',
where
_ hm
" ~2kf'
the indicated integrations in (11) and (12) have as a result,
{kry {drra)^ ^ Nhm It _^,
{e) {dx) € \a
X [e"- 1 +e''P(Vr7) - 2^^
+ «(5) (13)
SHOT NOISE IN DIODES
597
and
{kTY {drjffY- ^ Nhm
{e) {dx) €
\ a
X
e" - 1 - e'-PCVr?) + 2
+ /3(5), (14)
where
.=^(F'-F),
2 /"^
■rfx.
The fact that both a{b) and ^{b) are very small greatly simplifies
the solution for the distance coordinate x in (13) and (14). The
process is to invert the two equations, respectively, extract the square
root, and then expand the right-hand side in powers of a(6) and /3(5),
respectively. This results in expressions for dx/dr] which can be
integrated term by term. However, the small values of a(8) and /3(5)
allow powers higher than the first to be disregarded, and hence.
F(v')
1
kT'
and
«^ r Nhm
f(v)
3/2
I
"i' a(8)dr]
(15)
L
^-'Y
1. f
r Nhm lir , 1
(16)
where for convenience
r —
Jo r el - 1
C?T7
/(>?)
Jo
[
+ e^P(Vr,)-2^^j^'"'
Te"- 1 -6''P(V^) + 2^^r'
(2)
^o==^(F'- V,),
B =
/('Jo) =
C
Fir),') +/(77o) , dFir),') , ^/(r^o)
+
^lo'
dr]o'
dx
+
^TJc
eIodf(r]c
kT dr]o
e^ - 1 +
(Jil-iJJ'"
.[
if
1 - e-P(Vx) + 2
V^]"
"' [Vy" + X — y]
$(x)
(ix.
Vtt
[Vy- + X — y']d.
X
1 - e-P(^'x) + 2
3/2
$(x) = t^ - 1 + e^P(4x) - 2
P(x) =A r%-2^x.
Vtt Jo
Part II — General Discussion
The analysis in Part I shows that as soon as a potential minimum
exists, the tube noise in a planar diode is equivalent to the thermal
(45)
dxdydz
dxdydz \ , (46)
B
(47)
606 BELL SYSTEM TECHNICAL JOURNAL
noise of the plate resistance at an effective temperature which is a
function of that of the cathode.
In general, the effective value of the diode plate resistance tempera-
ture for any operating condition is very difficult to obtain because of
the complexity of the final noise equations (45) and (46). However,
the limiting value of the ratio of the effective plate resistance tempera-
ture to that of the cathode, denoted by "X" in (45), may be evaluated
very readily for certain limiting conditions.
One encounters the first of these conditions when the plate potential
and cathode emission are such that the potential minimum has moved
just up to the cathode, and is in fact on the point of disappearing.
This condition is secured by decreasing the space charge to values less
than are required for the formation of a potential minimum away from
the cathode. In the equations, it is represented by letting the quantity
770' approach zero, where 770' is the natural logarithm of the ratio of the
saturation current to the anode current. For this set of operating
conditions, all the electrons emitted from the cathode will go to the
anode, and hence the condition is appropriate to the study of pure shot
noise.
A second condition is obtained when the plate potential is equal in
value to the potential of the minimum. Physically, this condition
means that the minimum has moved just up to the anode, and requires
a negative value for the plate potential. Mathematically, it is repre-
sented by a zero value for the quantity 170, where 770 is equal to the
difference between 770' and (e/kT)Vp. For negative plate voltages
greater in magnitude than that of the potential minimum, all electrons
having an initial kinetic energy greater than eVp will reach the anode
regardless of the presence of the space charge existing between the two
electrodes. For these conditions, the diode becomes a temperature
limited current device.
A third limiting condition occurs when the plate potential is large in
magnitude compared with that of the potential minimum referred to
the cathode. In this condition a potential minimum still exists. It is
represented in the mathematics by letting the quantity 770 become
large. This condition represents the normal operating condition for
the diode.
As the space charge is decreased, making 770' very small, from (47),
the diode plate impedance becomes very large through the action of
dF(r]o')ldr]o' which becomes infinite as 770' approaches zero. As all
other quantities involved remain finite, the mean square noise current
for a very small space charge is
SHOT NOISE IN DIODES 607
dfM
ell} dr]o
Thus, as the potential minimum voltage is reduced to zero, the tube
noise as given by (45) reduces to the well known shot effect equation.
For some space charge at the cathode, the value of X in (45) has
definite limiting values for both very low and for very large plate
voltages. For a very small value of 170, that is for negative plate
voltage, the value of B defined in (47) is very large because rf/(»7o)M'7o
becomes infinite as rjo is decreased to zero. Thus as 170 —> 0
52 r y^-y-'dy = ^ ' (49)
Hence, for any value of space charge, the effective plate resistance
temperature for negative plate voltages is one-half of the cathode
temperature, under the restriction that no potential minimum exists
between the cathode and anode.
Since the diode is usually operated with a positive plate voltage, the
value of the effective plate resistance temperature for a large value of
770 is of more interest. For vo' not equal to zero, and a large value of
plate voltage, it can be readily shown that the values of /(ijo) and of D
are much larger than any other quantities involved in the equation for
X. After a bit of mathematical operation, it may be shown that the
limiting values for/(r7o) and D are
^ =W' [t""'" + ^^^'?«''' + . . . - {4yvo"' +•••)]
7^1/4 r 4 ,-
m2 L "^
From these relations, the limiting value of X for a large plate voltage
is given by
X = 3 J% r V23; - ^^ \~y'dy = 3(1-^) = 0.644. (50)
Thus, for any value of space charge, as long as a potential minimum
exists, a sufficiently large value of plate voltage may always be found
for which the effective plate resistance temperature is 0.644 times the
cathode temperature.
608 BELL SYSTEM TECHNICAL JOURNAL
It is possible to obtain a good approximation for the effective diode
temperature for any operating condition by the following method.
The values of "C" and "Z)" in (47) may be found without too much
difficulty by graphical integration for several different values of y, rjo
and rjo'. From the tabulated values for F(rjo') and /(rjo) given by
Langmuir, and from the values found for C and D, the integral,
S = I ylB - C - Dje~^'dy,
Jo
(5i:
may be evaluated by mechanical means for several values of t^o and
rio'. This gives the first integral in (46).
It was found practically impossible to calculate directly the contri-
bution to X from the last two integrals in (46). However, a rough
approximation to them may be found indirectly by the following
method : If the sum of the two integrals is denoted by Q, then (46) may
be written
^ dfivo) '
drio
or Q — B\(df(rio)/dr]o) — S = function of rjo' only.
For a fixed value of rjo', the solution of the above equation for several
values of r]o should give a constant value for Q. Unfortunately, only
the limiting values of X are known. However, if the limiting value of
0.644 is substituted for X in this equation, the calculated value of Q,
for a fixed value of r/o', should approach a constant value as t/o is
increased since X does assume the 0.644 value for t/o sufficiently large.
The limiting value of Q calculated in this manner is the desired
contribution to X from the last two integrals in (46). This method of
evaluating Q cannot be very accurate since it involves the difference of
two quantities of the same magnitude. However, since Q is small
compared to the contribution from the first integral in (46), a large
error in Q will introduce a much smaller error in the value of X.
The values of the effective diode plate resistance temperature
calculated in this manner for several different operating conditions are
shown in Fig. 2. These curves indicate that the effective diode
temperature is 0.644 times the cathode temperature for all practical
operating conditions. The values of 170' and 770 may be determined
from the following relations:
Tjo' = logey^, (52)
SHOT NOISE IN DIODES 609
where I^ is the saturation current and I p is the anode current, and
(53)
T?o ^ '^^' ~ yp ^^p ^ '^^' + "V" ^ 10"* ^p.
where Vp is the anode potential, and T is the absolute cathode temper-
ature.
For T = 900° K,
.70 = vo' + n.9Vp. (54)
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
—
0.2
^0 3
—
0.5
"^
wl/^'
^^S'
^ _
- "^
1
^=
=-
1
1.0
30
2.0
4 6 8 10 12 14 16 18 20 22 24 26 28 30
Fig. 2 — Effective noise generator voltage of planar diode.
£/- = ^krp{\T)df, rjo' = logey-', 570 = Va'
kT
Vr,
For T = 900° K., rjo = 770' + 12.9 Fp
Even for the small space charge condition for which the plate current
is eight-tenths of the saturation current (tjo' = 0.2), the value of 770 need
be greater than about 25 only before X assumes its limiting value. For
a temperature of 900° K., as for oxide coated cathodes, this would
require a plate voltage of only two volts. If the plate current were less
than eight-tenths of the saturation current for very high plate voltages,
then as the plate voltage is reduced, r/o' would increase. For this
operating condition X maintains its limiting value of 0.644 for all
except negative values of plate voltages.
The transition between the various effective planar diode plate
resistance temperatures is more clearly shown in Fig. 3. In this
610
BELL SYSTEM TECHNICAL JOURNAL
figure, the natural logarithm of the ratio of saturation current to the
plate current is plotted as a function of the plate voltage for several
constant values of the coef^cient X. These curves show for a fixed
positive value of plate voltage that as the space charge is decreased
toward zero, by a reduction in the ratio of the saturation current to the
plate current, the value of X for moderately large values of space charge
increases but little from 0.644, and then for a very low space charge,
increases very rapidly to its limiting value given by the shot noise
%
\
■ • ■
1
I
^o.6
1
>
. ' ■ •
^
I 1-0.644
I 1
11
m
1
>= 0.644
1
1
\
•.'•';•■•'•
•••Vi
■■'ill
1 \
1 \
V
^.
>v=0.5
0.675
V
s
'M{
1 /
--^
^^^
••'•!•*:
'M
f,^''
.—
0.8
-~:
-^.
.v.
•■"•':••
':iP\
0.9
— — .—
---
•~~i
~^c
-==z
^~~
-4 -2
20 22 24 26 28
Fig. 3- — Modification of effective plate resistance temperature produced by
space charge.
£/2 = ^krp{\T)dJ.
which is represented by the axis of abscissae. Thus, the value of X
digresses markedly from 0.644 only for the narrow region of operating
conditions for which the saturation current is less than 1.25 times the
plate current and the plate voltage is less than 2Se/kT volts. For an
oxide coated cathode for which T is 900° K., the effective plate re-
sistance temperature is 0.6447" for any operating condition for which
plate current is less than eight-tenths of the saturation current and the
plate voltage is greater than two volts.
For a cylindrical diode, the general method of analysis used in the
parallel plane case results in equations which are practically impossible
SHOT NOISE IN DIODES 611
to solve. The difificulty in these equations arises from the fact that
tangential as well as radial initial velocities must be considered in
obtaining the total anode current. Since it was shown for the planar
diode that the effective temperature of the plate resistance is 0.644
times the cathode temperature for practically all operating conditions,
all that is really desired in the cylindrical diode solution is the limiting
value of the effective tube temperature. This may be found rather
easily from a comparison of the cylindrical diode with the planar tube
in the following manner.
For a very large space charge, and a high plate potential the radius of
an equipotential surface near the potential minimum will be very
nearly equal to that of the cathode. Hence, for these operating
conditions, the planar diode equations may be applied to this region
of the cylindrical diode. In the planar tube, it was shown that for
770' > 3, 770 had to be of the order of unity to obtain the limiting value
of 0.644 for X. If the space charge and plate potential are sufficiently
large in the cylindrical diode, the radius of the equipotential surface
for which 170 is greater than unity will practically be equal to that of
the cathode. The cylindrical diode may then be divided into two
parts, a planar diode between the cathode and the equipotential
surface for which rjo > 1, and a cylindrical diode formed from the
remainder of the tube. In any diode, the only source of noise energy is
the cathode from which the noise power is transferred to the anode and
external circuit through the mechanism of the initial electronic
velocities. Furthermore, the same total noise power must be trans-
ferred across any equipotential surface between the cathode and anode.
In the planar portion of the cylindrical diode as described above, the
total noise power crossing any equipotential surface was shown to be
2.576kTdf. This same noise power must be transferred across any
other equipotential surface in the cylindrical diode. Hence, the
effective plate resistance temperature for the cylindrical electrode tube
must also be 0.644 times its cathode temperature. From this line of
reasoning, it may be shown that the limiting value of the effective
temperature for any shape diode is the same as that for the planar tube
with the same cathode temperature.
From the experimental data given in his paper, Pearson definitely
recognized that the limiting value of the diode plate resistance temper-
ature should be between 0.59 and 0.65 of that of the cathode.^ The
writer understands that North and Thompson of the R.C.A. in an
unpublished paper have obtained the same general result for the effect
of space charge upon shot noise in diodes.
612 BELL SYSTEM TECHNICAL JOURNAL
In a diode, the tube noise may be expressed equally well and with
equal correctness either as a modified shot noise or as a thermal
resistance noise. In this paper, the thermal resistance viewpoint was
taken for two reasons. First, the coefiicient "X," used in the thermal
resistance noise equation
£? = Akr,{\T)df,
is practically always a constant equal to 0.644, whereas, the factor,
"7^," used by Schottky and Spenke in their modified shot noise
equation
772 = 2eF'Iodf
is always a function of the operating condition. That is, for the
operating conditions for which X is a constant, F has the following
value:
1.39
[^''^Tr'^^T
The second reason for the selection of the thermal resistance noise
relation is that power from the motion of the atoms in the cathode is
actually transferred to the plate electrode and external circuit through
the mechanism of the initial electron velocities. Hence, the tube
noise in a diode with space charge is very similar to a thermal resistance
noise.
Part III — Effect of Transit Time
The analysis, in Part I, while giving the correct results for all
operating conditions in the ordinary frequency range, is extremely
long and cumbersome. It shows, however, that only the limiting
values of the effective temperature of the plate resistance are required
for most practical cases, and therefore it points the way to make
simplifying assumptions which result in a much shorter analysis, and
moreover, which allow the analysis to be extended to frequencies so
high that electron transit time phenomena become of importance.
Thus the final noise equation in Part I shows that for moderately
high anode potentials and for the usual excess of cathode emission,
a very good approximation may be had by a consideration of the
current-voltage relations existing in the jS-region between potential
minimum and anode without the necessity of encumbering the analysis
by including the a-region between potential minimum and cathode.
Moreover, for a large anode potential, the terminal velocities of the
SHOT NOISE IN DIODES 613
electrons at the plate are very large in comparison with their initial
velocities for practically all of the electrons. This means that the
transit time for the various electrons is practically the same for all
of them which leave the cathode within a particular very short time
interval, even though the initial velocities of the various electrons are
statistically distributed among them. It results that the various
individual velocities of the electrons in the /3-region may be replaced
by an average value, which at the potential minimum may be defined
as follows :
u'n{ii,)duc
-a--^ (55)
n{uc)duc
f
Physically, the meaning of this expression is the average velocity
of these electrons which cross a plane in the )3-region close to the po-
tential minimum in a unit of time. Inasmuch as the unit of time may
be taken to be very small, it follows that (55) expresses the effective
instantaneous value of the initial velocity which may, and does,
fluctuate as time goes on.
On the basis of an equation of the form
the planar diode has been extensively investigated by a number of
workers and it has been shown ^ that the relation between current
and voltage is completely specified as soon as two boundary conditions
are given. These may be the initial velocity and acceleration, or
they may equally well be the initial velocity and conduction current
pu. However, the analysis based on (56) applies strictly to the case
where all of the charge moves with the same velocity and hence
contains a certain approximation when electrons are considered whose
velocities have a certain dispersion around some mean value. The
error will be small until frequencies are considered which are so high
that a large proportion of the electrons which left the cathode in a
time interval which is very short compared with the period of the
high frequency arrive at the anode in a time interval which is not
small compared with the high frequency period. Normally this means
that the error is small even for frequencies so high that the majority
of the electrons require several cycles to make their transit from
potential minimum to anode.
614
BELL SYSTEM TECHNICAL JOURNAL
It is convenient to write the resulting equations in terms of d-c.
and first order a-c. values where the initial values of d-c. velocity
and acceleration are given, but initial values of a-c. velocity and
conduction current are employed. The first order a-c. relation derived
by Llewellyn may be written in the form
hm time nme
(57)
where qa and fXa are the initial values of fluctuation conduction current
and velocity, respectively, while A, B and C are defined by:
A =W - io^^x 4- -^ (2 ■- 2e-'"
id
11
B
C =
- -r^ [_aa{iee-^^ + e-'" - 1) + Uaio^ie-^' - 1)]
hmtisP'
(58)
in which ?? is the transit angle, wr, the transit time being r, and /o is
the d-c. current.
In the application of these relations to noise analysis, the initial
values of velocity, acceleration, and conduction current must be taken
at a point in the ^S-region beyond the potential minimum, but just as
close to it as possible without encountering conditions where electrons
may be moving toward the cathode, for the equations apply only to
cases where the electrons are moving in one direction only. The
initial point is, however, located so near to the potential minimum
that the d-c. acceleration in (58) may be taken as zero. When this
is the case, it may be shown that the initial conduction current is
equal to the total current. In other words, the initial value of dis-
placement current is zero. Under such conditions (57) and (58)
reduce to the following expression for the a-c. anode potential in
terms of the a-c. component of current and initial velocity :
V =
/i
nme \ 6
+ (^""Uaiie + e~^^ - 1)
-f '=^ [ide-'' + e-'o - 1]. (59)
co'e
The term multiplying the a-c. current h in the above equation is
the internal high-frequency impedance z of the planar diode. The
last^term may therefore be identified with an internal emf. When
the initial velocity /x, is expressed in terms of the fluctuation of electron
SHOT NOISE IN DIODES 615
velocity, the term gives the equivalent noise generator, E. Thus
E=f^{ide-^9 j^e-'9 - 1) (60)
and the mean-square value of the noise emf. (at a frequency w) is
given by :
£2 = ^^" \ie-^6 ^ Q-i0 _ u\
(61)
The problem is now reduced to finding the mean square value of
initial velocity fluctuation, txa^, which corresponds to electrons cross-
ing the potential minimum. This may be done by going to (55)
which gives the effective value of the instantaneous initial velocity
and separating all quantities, including the lower integration limits
into d-c. and a-c. components. Thus
n{uc) = no{tic) + 8(uc)
uj = Uc + 8Uc
u' = u^ -{- bu'
U = Ua + IJLa
(62)
The result may be expanded in series form and products of the 5's
may be disregarded inasmuch as the a-c. components are small in
comparison with the d-c. The indicated operations have as a result
and
e f"
Ma = -^ 1 (U' — Ua)8
{Uc)dUc
(63)
(64)
The Fourier analysis may be applied to this in the way outlined in
connection with (37) and (41) in Part I and gives the mean-square
value of velocity fluctuation corresponding to a frequency interval df
as follows :
— 2e2 p 4ekT,,/. 7r\
AT = T^«/ (" - Ua)-Uc{Uc)dUc = -f^^-dfi 1—1
1 0" ./_ I onm \ 4 /
(65)
This may be substituted in (62) giving for the effective noise emf. in
the frequency range df
E/ = 4kTdf\
el,
OT-
L hmt~
1
Xi[^- + 2 - 2 (cos d -\- 9 sin 6)']. (66)
616
BELL SYSTEM TECHNICAL JOURNAL
The initial average velocity is small so that the low-frequency
plate impedance may be written
ru =
eL
or"
12 //me'-'
(67)
Thus for any transit angle, the mean square noise generator voltage
is given by
£/ = 12 (^ 1 - -\kr,Tdf\j^[_2 + 0"- - 2(cos 6 + d sin ^)]
= ASk{OM^T)rpdf
S =
2 + e- - 2(cos d ^ 0 sin
(68)
For low transit angles, this expression reduces to
£.2 = 12 1
kr„Tdf,
(69)
which is precisely the limiting value obtained by the much longer,
but more rigorous analysis.
It must be understood that (68) is an approximation since the
transit time effect in the region between cathode and potential minimum
was entirely neglected, and because the validity of the average velocity
concept does fail at the very high frequencies.
Some knowledge of the extent of the operating conditions for which
the above equations are good approximations may be obtained from
the d-c. current-voltage relation. For the boundary conditions
assumed, the low frequency current equation derived from the general
solution given by Llewellyn reduces to
/ =
- 2.33(F
10'^(x — X
|^'[l+2.66^^-i^^]. (70)
This equation was shown by Langmuir to be a very good approxi-
mation for the plate current for most operating conditions and fails
only for very low values of plate voltages. Thus, it may be concluded
that (68) is a good approximation for all operating conditions except
for very low plate voltages and a small space charge.
The plot of (68) given in Fig. 4 shows that the magnitude of the
mean square noise generator voltage decreases by five per cent only
for transit angles as large as one radian.
SHOT NOISE IN DIODES
617
The effect of transit time on the pure shot noise for a low space
charge density and a high plate voltage may be obtained quite readily
from (57) and (58). Since for a very small space charge, Jo and Ua are
small, and a„ large, the equations then reduce to the following ex-
pression :
Vi = ^ — +-7^a„r-
1
- iiee~''> + e-
1)
(71)
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
^
\
\
\
\
\
\
\
\
s.
\
s^
^
n
0 1 23456789 10 II
TRANSIT ANGLE IN RADIANS
Fig. 4 — Effect of transit time on both thermal and shot tube noise.
E/ = 4SkiOM4:T)rpdf,
7/ = leSIodf.
For these operating conditions, the transit time in terms of the d-c.
acceleration and the electrode spacing is given by
X =
tta T-
In terms of the external circuit impedance Z/
(72)
618 BELL SYSTEM TECHNICAL JOURNAL
SO that (71) and (72) combine to give
h
qa
1 +
Zfioie
+ e-'" - 1;
(73:
The mean square shot noise current is thus given by
Ir =
1 +
Zfioie
I (i^
-\-e-
(74)
The value of the mean square a-c. conduction current at the cathode
to be substituted in the above equation may be derived as follows :
The total current emitted from the filament was defined as
J 100 /»00 /^OO
n(Uc)dUc = e I no(u,)diir + ^ I j d{Uc)duc. (75)
0 Jo Jo
Hence
5a
Jo
8(uc)dUc
(76)
From (37) and (41), the contribution to the mean square of this
conduction current from the frequencies between / and f -\- df is
ga~
..00
2e-dJ I «o(w,
Jo
)duc = lelodf.
(77)
With this result, the effect of transit time on the shot noise current is
given by
lehdf
1 +
Z /t(joe
-[2 + 6/2 - 2(cos ^ + ^ si
in0)]|.
(78)
where Z/ is the impedance of the external circuit at the frequency/ and
x/icoe is the capacitive reactance of the diode at the same frequency.
Thus the shot noise current is modified by transit time in precisely the
same manner as the noise generator voltage for the thermal tube
noise.
The effect of transit time upon the shot noise, as indicated in (78),
is identical with that obtained by Spenke for the same operating
condition of low space charge and high anode potential.* Spenke
derives this result through a clever application of a Fourier Series in
which account was taken of the effect of transit time upon the wave
shape of the current induced in the anode by the electron moving from
SHOT NOISE IN DIODES 619
cathode to the plate. The advantage of the method of average veloci-
ties used in this paper is that the effect of transit time in both the
thermal tube noise and the shot noise may be found.
It is noteworthy that Ballantine in 1928 derived an expression for
the effect of transit time upon the pure shot noise which is identical
to that obtained in this paper.^
In conclusion, the writer wishes to express his appreciation to F. B.
Llewellyn whose supervision and numerous suggestions made possible
this paper.
References
1. F. B. Llewellyn, "A Study of Noise in Vacuum Tubes and Attached Circuits,"
Proc. /.i?.£., vol. 18, pp. 243-265 (1930).
2. G. L. Pearson, "Shot Effect and Thermal Agitation in a Space Charge Limited
Current," Physics, vol. 6, pp. 6-9 (1935).
3. W. Schottky and E. Spenke, "Die Raumladungsschwachung des Schroteffektes,"
WissetischaflHche Veroff tntlichungen aus den Siemens-Werken, vol. 16, pp. 1-41,
Aug. 6, 1937.
4. E. Spenke, "Die Frequenzabhangigkiet der Schroteffektes," Wissenschaftliche
Veroffentlichungen aus den Siemens-Werken, vol. 16, pp. 127-136, Oct. 8,
1937.
5. L. H. Germer, "Distribution of Initial Velocities Among Thermonic Electrons,"
Phys. Rev., vol. 25, 795 (1925).
6. T. C. Fry, "Thermonic Current Between Parallel Plane Electrodes," Phys. Rev.,
vol. 17, 441 (1921).
7. L Langmuir, "Effect of Space Charge and Initial Velocities," Phys. Rev., vol. 21,
419 (1923).
8. F. B. Llewellyn, "Operation of Ultra High Frequency Vacuum Tubes," B.S.T.J.,
Oct. 1935.
9. S. Ballantine, " Schrot-Effect in High Frequency Circuits," J.F.I., vol. 206, 159
(Aug. 1928).
10. T. C. Fry, "The Theory of the Schroteffekt," J.F.I., vol. 199, No. 2 (Feb. 1925).
Fundamentals of Teletypewriters Used in the Bell System
By E. F. WATSON
During the past few years the use of teletypewriters has become
quite extensive in the Bell System. Simpler and cheaper machines
have recently been made available for meeting the simpler service
requirements and attachments have been designed to provide
additional features for meeting more complex service requirements.
This article discusses the fundamental principles and various
features of the teletypewriter machines now in common use and
explains the more important factors which have been controlling in
their development.
WITH the growth of Teletypewriter Exchange Service and the
general increase in the use of teletypewriters in private line
services of various types, questions frequently asked are: How do
teletypewriters operate? What is the "start-stop" system? Why is
it used? What is a regenerative repeater?
This article will attempt to answer some of these questions and
explain also the fundamental principles and features of teletype-
writers and their auxiliary arrangements as now employed in the Bell
System. These have been developed to meet the needs of customers
for a typed or similar record form of communication and at the same
time be suitable for operation in connection with the Bell System plant.
Code
For economical transmission over long distances it is fundamental
that only a single wire or transmission channel be required to carry
the signals. Furthermore, long experience with manual telegraphy on
land lines has proved that reliable and efficient operation is secured by
using not more than two conditions on the line, such as current and no
current or positive impulses and negative impulses, as contrasted with
the use of three or more conditions, or current values. The entire
telegraph plant of the Bell System as well as practically all other land
line telegraph systems have been built on this two-condition basis.
The familiar Morse code uses sequences of dots and dashes to
represent the different characters of the alphabet and meet the above
conditions. This code is not well adapted for teletypewriter control,
however, since the signals for different characters vary widely in the
time they require, from a single dot for the letter E to a combination of
620
FUNDAMENTALS OF TELETYPEWRITERS
621
several dots and dashes for some of the less frequently used letters or
numerals.
For machine operation it has thus far appeared desirable in order to
obtain simplest mechanisms and to obtain maximum operating speeds
with low line signaling frequencies, to have the signals for the different
characters of uniform length, that is, each contain the same number of
time units. This condition is met by the five-unit code where each
character is identified by the impulses in five units of time, and this is
the code normally employed in Bell System teletypewriters. Each of
the five units of this code may be either positive or negative, current
or no current, or either of two values of current, and the permutations
provided are 2'^ or 32. These are sufficient for the 26 letters of the
alphabet, a space, carriage return and paper feed signals as well as
case shifting signals to bring another set of characters into action so as
to include numerals and punctuation marks. A chart of this code as
used in Teletypewriter P^xchange Service (TWX), is shown below.
FIGURES
LETTERS
A
5
8
B
1
e
C
$
D
3
E
1
4
F
&
G
Q.
o
1-
ul
H
e
1
J
1
2
K
3
4
L
M
7
8
N
9
O
0
1
Q
4
R
_]
UJ
CD
S
5
T
7
U
3
8
V
2
W
/
X
6
Y
Z
Q
UJ
UJ
u.
UJ
z
-J
UJ
u
UJ
a:
cr
<
tr
UJ
UJ
_i
10
UJ
ce
u.
z
<
PULSE 1
• • • • • • •
• • • •••• ••
PULSE 2
• • ••••• •
• • ••• • ••
PULSE 3
• •••••••
• ••••• ••
PULSE 4
• • • • • •
PULSE S
• • • • • • •
• • ••••• ••
Fig. 1 — Chart of five-unit TWX code.
The keyboard used for sending this code is shown below.
Fig. 2 — Chart of TWX keyboard.
It will be noted that this keyboard is similar to the ordinary type-
writer keyboard except that there are only three rows of keys instead
of four as in the typewriter. In the typewriter keyboard, the lower
three rows of keys are used ordinarily for small letters but when a shift
622 BELL SYSTEM TECHNICAL JOURNAL
key is also operated they type the corresponding capital letters. The
fourth or top row of keys carries the numerals and certain punctuation
marks. The teletypewriter types capital letters but not small letters
so that by using a shift or Figures key the upper case position of the
letter keys is available for the usual punctuation marks and numerals.
Thus only three rows of keys are required on the teletypewriter key-
board. The operation of the Figures key sends a signal causing the
receiving machine to shift to upper case so that numerals and punctua-
tion marks will be printed until a Letters or Space signal is sent which
restores the machine to lower case.
Start-Stop System
For transmitting the signals of the five-unit code over a telegraph
line, it is necessary to have some system of timing so that each of the
five impulses may be properly received, identified and interpreted at
each receiving station. The start-stop system is used for this purpose.
One arrangement of this system using segmented distributors with
revolving brushes, is illustrated in Fig. 3.
In this system both sending and receiving brush arms are normally
at rest but are maintained under constant torque, tending to rotate
them in the direction of the arrows, by constantly running motors
driving them through friction clutches. Normally the line circuit is
closed and carries current. When a key of the keyboard is operated
to send a signal, the start magnet of the sending distributor is ener-
gised releasing the sending brush arm and allowing it to rotate. As
this brush passes from the stop to the start segment, the line circuit
is opened and this open signal transmitted to the receiving station
where it causes energization of a start magnet which releases the
receiving brush and allows it to rotate.
Both sending and receiving brush arms rotate at approximately the
same speeds since they are driven from motors running at approxi-
mately the same speeds. These motors are either small synchronous
motors driven from constant frequency commercial 60-cycle 110-volt
power supply or by commutator type motors, equipped with centrifugal
governors to hold them at approximately a constant speed, for use on
other commercial a-c. or d-c. supplies.
Now as the sending brush arm sweeps over the sending face, the
impulses of the five-unit code, as set up by the particular key de-
pressed, will be transmitted over the line as shown in Fig. 4 for the letter
A, and through the action of the rotating receiving brush, Nos. 1 and 2
current impulses will cause the energization of Nos. 1 and 2 selecting
FUNDAMENTALS OF TELETYPEWRITERS
623
0^\V:\V^-v^: \ 5.9 "/a SLOWER THAN
) RECEIVING DISTRIBUTOR
Fig. 5 — Effect of distorted signals on reception.
experienced in service without failure to receive and properly identify
each pulse as a No. 1, 2, 3, 4 or 5 pulse. The other traces illustrate
certain types of distortions which may be experienced and the condi-
tions existing for their proper reception with this adjustment. These
traces will now be explained on a purely theoretical basis assuming an
ideal receiving machine without mechanical or other imperfections.
Trace (b) shows the conditions in case of 25 per cent "marking bias"
in the received signals; that is, each marking pulse has been lengthened
FUNDAMENTALS OF TELETYPEWRITERS 627
by 25 per cent of one pulse length. It will be noted that under these
conditions each pulse will still be properly received and identified.
Trace (c) illustrates 50 per cent marking bias in the received signal,
and at this point it will be noted that the No, 3 and stop pulses have
been so elongated that they are just on the verge of being erroneously
received and identified also as No. 2 and No. 5 pulses. This then is a
theoretical limit for proper operation with marking bias without a
readjustment of the receiving distributor.
Similarly traces (d) and (e) illustrate respectively the conditions
when the received signals have 25 per cent and 50 per cent "spacing
bias," that is each marking impulse has been shortened by this per-
centage of one pulse length. It will be noted that 25 per cent spacing
bias can be easily tolerated but that with 50 per cent spacing bias the
No. 1 and No. 3 pulses are on the verge of failure to be recorded. This
then is a theoretical limit for spacing bias in the signals under this
adjustment.
Traces (f) and (g) show the effect of 25 per cent marking and spacing
distortions respectively on the rear end of the stop or front end of the
start impulse, all other pulses remaining undistorted.
Trace (h) shows the effect of distortions on the selecting impulses
alone. By combining this trace with traces (f) and (g) it will be seen
that with 25 per cent distortion of the start pulse, 25 per cent distortion
of the same sign is the limit for distortion on the front end of marking
impulses, or 25 per cent distortion of opposite sign on the rear end of
marking impulses. Thus for distortions other than bias, which are
apt to affect both start and selecting impulses in the same signal,
± 25 per cent is the theoretical limit of allowable distortion.
Traces (i) and (j) show the effects of speed inaccuracies. From
these it will be seen that theoretically the sending distributor could be
about 8.9 per cent faster or 9.5 per cent slower than the receiving
distributor before errors would be experienced.
In practical machines there are, of course, inaccuracies due to
tolerances of manufacture, and other departures from the ideal so that
the above mentioned theoretical limits are not reached. However,
all machines used in the Bell System are required to tolerate a lengthen-
ing or shortening of the front end of any current impulse of at least
40 per cent of its length and with the same adjustment a lengthening
or shortening of the rear portion of any current impulse of at least 35
per cent with the start pulse undistorted. Since bias is nearly always
present to some degree in the received signals, and since as interpreted
by the receiving distributor it affects only the front end of the current
impulses as illustrated in traces (b), (c), (d) and (e) of Fig. 5, the
628
BELL SYSTEM TECHNICAL JOURNAL
distributors are usually adjusted for maximum tolerance of front end
distortions and then have ample tolerance for such distortions of the
rear ends of the current impulses as are experienced under service
conditions.
Regenerative Repeaters
As circuits become longer and more complex, eventually a point
is reached beyond which signal distortion becomes so great that the
signals cannot be reliably received without error. To overcome this
Fig. 6 — Regenerator unit.
limitation a device known as a regenerative repeater may be inserted
in the line at this point. It has a receiving mechanism similar in
principle to the receiving distributor of a teletypewriter and will
accurately receive and interpret any signals which a teletypewriter
would accurately record. This receiving mechanism is interconnected
with a retransmitting mechanism, or sending distributor, which re-
transmits the signals reshaped and reformed so as to be substantially
free from distortion. In its latest form a one-way regenerative re-
peater consists of a receiving magnet and a set of transmitting contacts
interconnected by some relatively simple mechanical parts driven from
FUNDAMENTALS OF TELETYPEWRITERS 629
a motor. A photograph of such a recently developed regenerative
repeater unit is reproduced in Fig. 6.
By means of these regenerative repeaters reliable teletypewriter
service may be extended to any desired distances so long as the signals
in any one regenerative repeater section are not too badly distorted to
permit reliable operation of that section alone. Several regenerative
repeaters may be operated in tandem on a single very long circuit if
required and in fact a number of very difficult long circuits are operat-
ing satisfactorily under these conditions at the present time. A point
worthy of note and which has not previously been mentioned is that
the stop impulse in the code adopted for Bell System apparatus is
slightly longer than the other impulses, which facilitates the use of
regenerative repeaters in tandem without requiring complex speed
control arrangements.
General Features of Teletypewriters
Teletypewriters are widely used for high speed written communi-
cations. Generally speaking, written communications are desired for
purposes of accuracy. Therefore, high speed, accuracy and reliability
are basic requirements for teletypewriter service.
In choosing an operating speed at which distributors of teletype-
writers are to be set, several factors must be considered. These are the
capabilities of the mechanisms of the machines, the average capabilities
of operators for continuous sending at high speeds, the commercial
need for high speeds, and the capabilities of the line circuits for
transmitting the signals reliably over long periods without excessive
distortions or excessive attention for maintenance and adjustment.
A satisfactory compromise among these different factors seems at the
present time to be about 60 words per minute, or 368 machine opera-
tions per minute, which is the speed usually employed in the Bell
System. The machines themselves may be arranged and adjusted
to be capable of higher speeds up to about 75 words per minute, and it
may be that, in the future, service at these higher speeds will be
justified under certain circumstances.
Accuracy and freedom from breakdown troubles are necessarily
inter-related and both required to a very high degree for machines
handling important written communications over long distances. To
give some idea of the severity of these requirements, we have found
from long experience that to produce a good machine we can not be
satisfied in our laboratory tests unless the machine is capable of typing
at least 1,000,000 consecutive words (6,000,000 operations) without
630 BELL SYSTEM TECHNICAL JOURNAL
error or trouble of any kind, and without requiring service attention
of any kind other than normal replacement of paper and inking
ribbons.
For rendering service economically with teletypewriters on sub-
scribers' premises, an important requirement in order that the expense
of maintenance be not prohibitive is that the machine should not
require maintenance attention except at very infrequent intervals.
Bell System machines are designed to require routine maintenance
attention not oftener than once in two months where the machine is
used continuously over periods of eight hours each day. To accomplish
this the problem of lubrication has required very careful attention. It
has necessitated the provision of oil reservoirs in certain places and the
careful selection and specification of oils and greases. Another feature
making for economical ma.n c : nance is interchangeable parts. In other
words, if a part breaks or wears, it is replaceable by another part of the
same type without requiring fitting and usually without readjustment.
At times customers wish to use teletypewriters on tables especially
designed and arranged to suit the convenience of their offices. For
this reason teletypewriters are designed as far as feasible to be self-
contained units which can be mounted on any desk or table.
All present Bell System teletypewriters employ the start-stop system
of synchronizing and are well adapted for the connection of any
number of machines to one circuit with facilities for rapid to and fro
intercommunication among the various stations. To permit optimum
control of intercommunication and interruption of the sending station
when desired, a device known as the "break lock" is incorporated in
many machines. This device, together with a "break" key located on
each machine, provides facilities whereby any station may interrupt a
station which is sending, take control of the circuit and send. The
operation of the "break" key opens the line transmitting a signal which
causes the "break lock" device to function at the station which is
sending and automatically stop any further sending from that station
until the device is manually restored. This device is very important
in the case of transmission from a perforated tape, which is described
later.
Motor control devices are of importance for stations which are not
in continuous use but which may wish to receive messages from time
to time from distant stations without requiring an attendant to turn on
the machine. Such devices are used both in private line and in TWX
services. In the case of a private line it is often desired to have the
machine normally idle with the motor stopped but so arranged that.
FUNDAMENTALS OF TELETYPEWRITERS 631
when a distant station wishes to send a message, a signal may be sent
which will automatically start the motor and condition the receiving
machine so that it will properly record the message and then have its
motor automatically stopped again at the end of the communication.
Various devices are available for this purpose, some operating over
the regular signaling circuit and others requiring a separate circuit.
Similarly, in the case of TWX service, stations may, if desired, be
equipped for unattended service so that, if the station is called and no
attendant is present, the teletypewriter motor may be started remotely
by the switchboard operator and the station conditioned to record the
incoming message at the termination of which the motor can again be
stopped by the switchboard operator.
Signal bells are usually provided on the machines so that, if it is
desired to call an attendant to a working machine or to call attention
to a specially important message being received, the bell can be rung
by signals sent over the circuit.
A general feature incorporated in the design of all modern machines,
and one which is not often appreciated, is the so-called "overlap."
This feature makes high speed possible by overlapping the selecting
and printing parts of the receiving operation. In other words it
provides for the typing of one character to take place simultaneously
with the reception of the selecting impulses for the next character.
Features of Page Teletypewriters
Page teletypewriters have been built in several different forms,
notably with a moving paper carriage or a stationary paper carriage and
with a typewheel or with type bars for printing. An early design
employed a moving paper carriage and a typewheel, with an ink roller
for inking the characters on the wheel. With this design it was im-
practical to make satisfactory carbon copies, the printed record
was unevenly inked, and much trouble was experienced due to side
printing, that is, unwanted printing of portions of letters adjacent to
the desired letter on the typewheel. Furthermore, considerable
trouble was had in properly feeding paper from a paper roll through the
moving paper carriage.
To eliminate these limitations and troubles it was decided that for
general service in the Bell System a new machine should be designed to
be capable of making as many carbon copies as a typewriter and that
it should use type bars and have a stationary paper carriage. This
sort of machine was new in the art and required extensive development
work to produce a satisfactory commercial design because of the in-
632 BELL SYSTEM TECHNICAL JOURNAL
herent difficulties of, moving an automatically operated basket of
typebars back and forth in front of the stationary paper. The present
standard No. 15 teletypewriter was the ultimate result of this work
and has proved very satisfactory in general service over a number of
years. It employs a typewriter ribbon for inking, has the paper roll
inside the machine cover and makes very satisfactory carbon copies
with various types of paper supply without being subject to the paper
feed, inking and side print troubles previously experienced.
This machine has also lent itself to meeting later demands from
business houses for typing either single or duplicate copies on special
printed forms as commonly used in modern business practice. By
equipping the platen with sprocket teeth and having feeding perfora-
tions along the edges of the forms, all copies of these forms are auto-
matically held in perfect registration during typing at all stations
connected to the circuit. In connection with the rapid handling of
these forms a further requirement for automatic tabulation has been
met by providing a tabulating device which on the transmission of a
certain signal causes all carriages to move over rapidly to any pre-
determined position on the form and stop there for the typing of letters
or figures in columns perfectly aligned. This device greatly facilitates
the rapid transmission and reproduction of orders and the like on
organized printed forms.
With the advent of TWX service a new situation arose in which many
of the machines were only infrequently used and then for very short
periods to make a single copy only. To render this service economi-
cally it seemed desirable to have a less expensive machine and since
narrower capabilities were required this seemed entirely feasible.
Accordingly a new machine known as the No. 26 teletypewriter has
been developed primarily to print a single satisfactory copy although
one carbon copy can be made if desired. To obtain low first cost this
machine has a moving paper carriage and to secure a satisfactory
printed record it employs ribbon inking and a typewheel arrangement
which is a sort of cross between conventional typebar and typewheel
designs. This typewheel is an assembly employing a small individual
type pallet for each separate character. In the process of printing a
character, a striking arm somewhat like the shank of a typebar comes
forward and forces the individual type pallet against the ribbon to
make an impression on the paper. The typewheel is rotated to
different positions to select the different characters to be typed. In
this way satisfactory inking and a clear cut impression without side
print is obtained, which compares favorably with the record obtained
FUNDAMENTALS OF TELETYPEWRITERS
633
on a typebar machine or typewriter. The entire machine costs
appreciably less than the more comprehensive No. 15 machine. The
No. 26 machine is illustrated in Fig. 7.
Fig. 7 — No. 26 teletypewriter.
Features of Tape Teletypewriters
In the case of tape teletypewriters it is also necessary to have a clean
printed record and occasionally there is a need for carbon copies.
Accordingly, the tape machine standard for the Bell System is a type-
bar machine using an inking ribbon and known as the No. 14 teletype-
writer. It is illustrated in Fig. 8.
634
BELL SYSTEM TECHNICAL JOURNAL
A feature worthy of note is that with this machine typing always
occurs at the same point introducing a problem in connection with
platen wear. If the platen were fed by the usual ratchet in, say, 36
steps per revolution, there would be heavy wear concentrated at these
36 points and the platen would require frequent replacement to pre-
Fig. 8— No. 14 teletypewriter.
serve good printing. To avoid this, the platen is fed through differ-
ential gearing so that on a second revolution the typing comes in a
different spot from that of the first revolution; thus the wear is uni-
formly distributed over the entire circumference.
One carbon copy can be made by leading tapes through the machine
from two rolls of record paper and one roll of carbon paper. Two
carbon copies can be made in a similar way if desired.
FUNDAMENTALS OF TELETYPEWRITERS 635
The tapes employed may be either gummed on the back for con-
venient pasting on blanks for filing or may be plain paper tapes if the
records are of temporary interest only. Also cellophane or similar
transparent tape may be used if it is desired to project the record on a
screen. A tape out signal is provided on the machine so that when a
roll of tape becomes nearly exhausted a bell will ring continuously to
give warning of this fact. Where a bell is not desired, the last few feet
of tape on the roll are painted red to give similar warning.
If desired, this tape printing machine may be used on the same
circuit with page printing machines such as the No. 15 teletypewriter,
and when so employed is usually equipped with an "end of line
indicator" to warn the operator of the approach of the end of the line
in the page machine, so that suitable signals may be sent for starting
a new line.
Features of TWX Switchboard Operators' Teletypewriters
Such machines must be small in size to permit their use in a switch-
board position, quiet in operation to permit their use in the same room
with a telephone switchboard and must be capable of working with
any machine employed in the TWX system.
To meet these requirements the standard No. 14 tape teletypewriter
has been modified in several important respects as follows:
1. It has been provided with a specially designed enclosing cover
which reduces the machine noise radiated by at least 5 db more than
standard covers.
2. The machine is tilted so as to raise the keyboard and permit the
operator to assume a more elevated position nearer the switchboard
jack field.
3. It is equippent with an end of line indicator mechanism and lamp
to warn of the approach of the end of a line when sending to a page
teletypewriter station so that the proper signals may be sent to start
a new line.
4. The usual tape feeding mechanism which pulls the tape past the
typing point and obscures some of the typed message is replaced by a
so-called "push feed" mechanism which acts ahead of the typing point
and makes the typed message more fully visible.
5. Many of the operators' machines are provided with specially
arranged power supply and governing circuits so that their motors
normally run from 115 volt a-c. commercial supply but in case of a
power failure can be quickly switched to run from the 130 volt d-c.
telegraph battery.
636 BELL SYSTEM TECHNICAL JOURNAL
Features of Monitoring Teletypewriters
In connection with private wire teletypewriter service it has been
found very desirable to have so-called monitoring teletypewriters in
the repeater offices to facilitate testing between offices and with the
subscriber stations. These machines must be adaptable to work with
any subscriber's machine and to be usable for making test measure-
ments on circuits.
The No. 14 tape teletypewriter has also been adapted to this service.
It may be equipped with an end of line indicator to facilitate communi-
cation with a page teletypewriter. Also since commercial service is
given at speeds of 40 and 60 words per minute, many of the monitoring
machines are equipped with two-speed governors and a switch to
provide for changing from one speed to the other. These machines
are also usually arranged for normal operation from commercial power
supply but emergency operation from the 130- volt telegraph battery.
For making test measurements over circuits a special orientation
scale is provided together with a small crank extending through the
cover for quickly shifting the orientation setting to any desired point.
With the machine carefully adjusted to be practically free of harmful
distorting effects on the signals, it may then be used for measuring
distortions in received teletypewriter signals, the scale being arranged
to read the total distortion directly in percent of a pulse length.
Tape Storage Transmission
A heavy volume of traffic may be transmitted rapidly and con-
veniently by the use of perforated tape. In this method a machine
known as a perforator and having a keyboard like that of the teletype-
writer is used for punching the code signals for the message in a strip
of paper tape. This may be done with simultaneous typing of the
message on the teletypewriter in which case the speed of perforating
is limited to the speed for which the teletypewriter is set. If a typed
record is not made simultaneously with the perforating, punched tape
may be prepared at practically any speed within the capabilities of the
operator. This punched tape may then be fed through a device known
as a tape transmitter which automatically transmits the message
signals from the tape at the maximum speed for which the teletype-
writers connected to the circuit are set, which is usually 60 words per
minute.
The method of transmitting from perforated tape has the distinct
advantage of using the line at maximum efficiency at all times as com-
pared with direct keyboard sending where pauses in operating the keys
FUNDAMENTALS OF TELETYPEWRITERS 637
and interruptions to the operator result in direct losses of circuit time
and effectively slower transmission.
Another important advantage of the perforated tape method is that
errors may be corrected in the tape before transmission with the result
that only errorless copy is transmitted on the circuit. This is done in
the following manner and is illustrated in the section of perforated
tape shown below. If the operator in attempting to write the word
THE should strike the keys T and J (in error), realizing her error she
back spaces the tape one division, strikes the "letters" key and then
the correct keys H and E. The transmission of the "letters" signal
will cause no operation in the recording teletypewriters since they are
already in the "letters" case, and the word will be recorded correctly
as though no error had been made. Similarly, entire words or groups
of characters may be erased from the tape if desired.
LETTERS FEED
SPACE — ^ I ^ SPACE HOLES
• • • • •
• • ••••
• •• •• •
•••• • ••
•• ••• • •
FROM T HE LARGE
Fig. 9 — Sample of strip of perforated tape.
For TWX service a further advantage of the perforated tape method
is that the entire message may be punched in tape and checked by
printing, if desired, before a call is placed and a connection established.
Then, when the connection is established, the message can be auto-
matically transmitted at maximum speed requiring a minimum time
for the connection and giving a minimum charge for the call.
It is true, of course, that in this method there is some delay between
perforation and transmission. For this reason short to and fro mes-
sages, as required in setting up a connection, may be better handled
by direct keyboard. To facilitate such working, the perforator key-
board is normally arranged so that by throwing a switch this same
keyboard may be used for direct keyboard sending without perforating.
This switch also has an intermediate position in which the keyboard is
connected for simultaneous direct sending and perforating. This
provides for meeting the needs of certain TWX subscribers who wish
to simultaneously type and punch the message so that the typed copy
may be checked as it is perforated.
The complete page printing set arranged for tape transmission is
638
BELL SYSTEM TECHNICAL JOURNAL
known as the No. 19 teletypewriter set and is illustrated in Fig. 10.
It employs a No. 15 teletypewriter as the page printing unit.
Fig. 10 — No. 19 teletypewriter set.
Automatic Retransmission Using Reperforators
At times it is desirable to retransmit messages received from one
circuit to some other machine or machines on a separate circuit. A
unit known as a "reperforator" is often used to facilitate such re-
transmission. The reperforator now standard for the Bell System is a
start-stop receiving device using the 5-unit permutation code. It is
somewhat similar to the receiving-only tape teletypewriter except
that the record produced consists of code perforations in a tape rather
FUNDAMENTALS OF TELETYPEWRITERS 639
than typing on a tape. This perforated tape is the same as tape pro-
duced by a keyboard perforator as previously described, and may be
used in an automatic transmitter for retransmitting the message on a
separate circuit. The reperforator is usually associated with a re-
ceiving teletypewriter on a circuit and may be cut in or out manually
or automatically from signals transmitted along with the message
signals, so that it will automatically reproduce a code tape for use in
automatically retransmitting such messages as desired on some new
connection.
Conclusion
The fundamentals of teletypewriters, as described above, now seem
to be fairly well established. The future should bring simpler and
cheaper machines, especially where the more difficult requirements
do not have to be met, and probably additional attachments and
auxiliary features to extend the applications and convenience of
operation.
The Dielectric Properties of Insulating Materials
By E. J. MURPHY and S. O. MORGAN
This article discusses the variation of dielectric constant and
dielectric loss in the radio and power frequency range with the
object of giving a simple picture of the type of mechanism which
is able to produce anomalous dispersion in this range of frequen-
cies. Some of the general characteristics of anomalous dispersion
can be demonstrated as well on a simple and arbitrary model of the
structure of dielectrics as on the more complex ones which corre-
spond more closely to the actual structure of dielectrics. Such a
derivation is given here in order to indicate the significance of the
different factors which occur in the formulee which have been
proposed to account for the variation of dielectric constant and
dielectric loss with frequency. This enables a distinction to be
made conveniently between the general characteristics which are
shared by several types of dielectric polarization and the special
characteristics which are peculiar to a restricted class of polariza-
tions or to a particular kind of polarization.
II. Dielectric Polarizability and Anomalous Dispersion
IN a previous paper ^ the general features of the dependence of
dielectric constant on frequency were indicated schematically for
the entire range extending from the frequencies used in power trans-
mission to those of ultra-violet light. In the range of frequencies
below the infra-red (that is, in the electrical range of frequencies)
anomalous dispersion is the rule, normal dispersion not having been
observed as yet, except for piezo-electric materials, whereas at high
optical frequencies normal dispersion is the predominant feature. In
the intermediate infra-red region it is not surprising to find a behavior
which shows anomalous and normal dispersion in more nearly equal
degrees of prominence.
It will be recalled that anomalous dispersion is the type of frequency-
variation in which the dielectric constant decreases with increasing
frequency, while normal dispersion is the reverse of this, the dielectric
constant or refractive index increasing as the frequency increases.
The use of the term anomalous dispersion to describe the dependence
of dielectric constant on frequency in the radio and power frequency
range is now widespread, and seems quite appropriate, for it brings out
the point that the variation of dielectric constant with frequency in
» Murphy and Morgan, B. S. T. /., 16, 493 (1937).
640
DIELECTRIC PROPERTIES OF INSULATING MATERIALS 641
the radio and power range is in certain respects the same type of
phenomenon as optical anomalous dispersion.
Anomalous dispersion plays a very important part in the behavior
of dielectrics in the electrical range of frequencies. It is seldom possi-
ble to interpret a set of measurements of dielectric constant or other
dielectric properties without encountering some manifestation of
anomalous dispersion or of the other characteristic types of behavior
which follow as corollaries of it.
The two catagories, polarizability and dispersion, include a great
deal of the dielectric behavior of insulating materials. This paper
will deal primarily with anomalous dispersion, but the theory of anom-
alous dispersion is not entirely separable from that of the polarizations
of which it is an attribute, so it will be necessary to discuss at least
briefly the nature of dielectric polarization.
The Relation between Polarizability and Dielectric Constant
For our purposes a dielectric may be thought of as an assemblage of
bound charges, where this term is intended to include the electrons and
positive cores in atoms and molecules, the ions held at lattice points in
ionic crystals and, in general, any assemblage of charged particles which
are so bound together that they are not able to drift from one electrode
to the other under the action of an applied electric field of uniform
intensity. Actual dielectrics, of course, also contain some conduction
electrons or ions which are free to drift through the material and dis-
charge at the electrodes, producing a direct current conductivity.
This conductivity is small at ordinary temperatures in materials
classified as dielectrics.
The positions of these charged particles may be considered to be
determined by an equilibrium of forces. When an electric field is
applied this equilibrium is disturbed and the bound charges are dis-
placed to new positions of equilibrium; then when the applied field is
removed they revert to their initial positions. In the equilibrium
positions which the charges occupy when a constant electric field has
been impressed on the dielectric they have a larger potential energy
than in their initial positions. Moreover, they do not revert instantly
to their initial positions, and when the retardation is due to friction
some of the potential energy of the bound charges is dissipated as
heat in the dielectric.
When an alternating voltage is applied to the dielectric, we may think
of the bound charges as moving back and forth with certain amplitudes,
a different amplitude for each different type of bound charge. When
the applied electric field is of unit intensity, the sum of the product of
642 BELL SYSTEM TECHNICAL JOURNAL
amplitude and charge extended over all of the bound charges in a unit
volume of the material determines the dielectric constant of the mate-
rial. The energy dissipated as heat by the motions of these bound
charges in the applied electric field represents the dielectric loss per
second, a quantity which is proportional to the a.-c. conductivity
after the d.-c. conductivity has been subtracted from it. The ima-
ginary part of the complex dielectric constant is proportional to the
dielectric loss per cycle.
While the physical meaning of the dielectric constant and dielectric
loss can be conveniently described, as above, in terms of the amplitudes
and energy relationships of bound charges in their motions in an
applied electric field, a more useful basis for the discussion is that
provided by the concept of polarizability. In the present application
the polarizability is equivalent to the product of charge and amplitude,
but it has the advantage of being a quantity which is defined and dis-
cussed in the general theory of electricity as well as in that of dielec-
trics. The dielectric constant is then found to be related closely to
the polarizabilities of the assemblages of charged particles which the
dielectric contains.
The polarization of an assemblage of charges is a quantity defined in
electrostatic theory as the vector sum
P = HCiSi, (1)
where Si is the distance of the i^^ charge, d, from a point chosen as
origin, and the summation is extended over all of the charges in the
assemblage, for which Ci is a typical charge. (If the assemblage has no
net charge (X^i = 0)- the origin may be arbitrarily located without
affecting the value of p.)
The polarization is a vector quantity. It can be written as the
product of a scalar quantity p, which represents the magnitude or
electric moment of the polarization and a unit vector pi which gives
the direction of the polarization ; thus p = ^pi. As it will not be neces-
sary to distinguish between the properties of isotropic and anisotropic
materials in this article the direction of the polarization need not be
emphasized. The notation will therefore be simplified, in general, by
using the magnitude or scalar part of such vector quantities as the
polarization, the electric field intensity and the displacement of charged
particles.
To illustrate the application of equation (1) let us consider a very
simple configuration consisting of two charges + e and — e (see Fig. 1),
The vector polarization of this configuration is p = e(si — S2) = ^pi,
where p is the magnitude or electric moment of the polarization and
DIELECTRIC PROPERTIES OF INSULATING MATERIALS 643
pi is a unit vector in the direction of the vector (si — S2). If now one
of these charges is an electron {e — 4.77 X 10"^° e.s.u.) and the other
a unit positive charge and they are separated by a distance of the order
of magnitude of atomic distances (10~^ cm.), p has the value 4.77
X 10~^^ e.s.u., or 4.77 Debye units. The permanent electric moments
of molecules seldom exceed a few Debye units.
Let us now apply the definition contained in equation (1) to a
dielectric material. In the first place it indicates that if we know the
effective positions of the electrons and other charged particles which
Fig. 1 — The calculation of the polarization vector by the general method for a verj'
simple configuration.
contribute to the structure of the material we can always, in principle,
calculate the polarization of the body as a whole or any part of it.
Actually the calculation of the polarization of a body as a whole or
that of unit volume in it is in general a complicated matter involving
statistical considerations, but there are special cases in which the result
is rather obvious. For example, in a gas or liquid if all orientations of
the molecules are equally probable in the absence of an applied field,
the value obtained by taking the time-average of the summation indi-
cated by (1) is zero. Equation (1) would also give the value zero when
applied to all of the ions in a c.c. of a solution because any arbitrarily
chosen small volume in the liquid would be as likely to contain a posi-
tive ion as a negative ion.
644
BELL SYSTEM TECHNICAL JOURNAL
In some crystalline materials equation (1) gives the value zero be-
cause there is a suitable symmetry in the configuration of charged
particles in the unit cell ; for other solids equation (1) gives a finite value
for the unit cell, but zero when applied to a volume of the material
large enough to contain a great many crystallites with random orienta-
tions; however, there are some macroscopic crystals which have per-
manent polarizations. A solid material consisting of polar crystallites
with random orientations is analogous, as far as equation (1) is con-
cerned, to a liquid or gas containing polar molecules having random
orientations; the polarization of the material as a whole is zero in
either case.
CONDENSER-
PLATES
• POSITIVE CHARGE
O NEGATIVE CHARGE
Fig. 2— A dielectric in a condenser. The circles joined by a bar represent "bound
charges" of various kinds, including atoms and molecules.
Let us now consider a dielectric of any kind occupying the space
between two plane, parallel condenser plates of great enough area and
small enough separation that the electric field between the plates when
they are charged may be considered to be directed normally to them
(cf. Fig. 2). Consider the space between the plates of the condenser
to be divided into small" cubes of the same size, the purpose of this
imaginary division of the dielectric being merely to obtain a representa-
tive specimen of the dielectric material. If the cube size is too small
the instantaneous value of p obtained by applying equation (1) to all
of the particles in a cube will vary appreciably from one cube to another ;
DIELECTRIC PROPERTIES OF INSULATING MATERIALS 645
but we can then increase the size of the cubes until p is the same for
each cube to a close enough approximation. The polarization in each
cube is then representative of that of the dielectric as a whole, ^ and by
dividing X! ^i Si for a typical cube by the volume of the cube we obtain
the polarization per unit volume, which for the present will be designated
as P. This quantity is a statistical mean value involving a summation
over a large number of particles; its value depends not only on the
structure of the material but upon the effect of thermal motions on the
mean positions and orientations of the molecules or other elementary
particles in the material. One of the most interesting points in dielec-
tric theory is the consideration — pointed out by Debye and at the
basis of his theory of polar molecules — that for some types of structure
the mean positions of the particles from which P is calculated are
unaffected by changes in the amplitude of thermal motions while for
another type of structure (consisting of polar molecules free to assume
many or at least several orientations) an increase of temperature de-
creases P, because the randomness of the orientations of the polar
molecules is increased.
For many materials P is zero when no electric field is applied, and
assumes a finite value only when an electric field is applied, though as
has been indicated, some crystalline materials have a finite value of P
even in the absence of an applied electric field. In either case, how-
ever, the application of an electric field causes the bound charges
within the dielectric to be shifted in general to new equilibrium posi-
tions, corresponding to the slight change in the system of forces acting
upon them, and if the material did not have a polarization before the
application of the field, it assumes one; if it did, it assumes a different
value of P. The value of P when an electric field E is applied will be
designated as Pe, and that when no field is applied by Pq. Then
Pe — Po is the polarization per unit volume induced by an applied
field £. As the dielectric constant of a material depends upon the
magnitude of the polarization induced in it by an applied field, and we
are concerned here with dielectric constants, it will be desirable to
simplify the notation by setting Pe — Po = P- This gives P a
slightly different meaning than it had in the earlier part of the dis-
cussion, where it represented the total polarization per unit volume
whatever its origin.
2 A detailed consideration of the method of dividing a dielectric up into ele-
mentary volumes in order to compute the mean polarization encounters complications
which need not be discussed here. A critical analysis of the method of computing
the volume density of polarization of a dielectric is given by Mason and Weaver,
"The Electromagnetic Field," Chicago (1929); Chapter III.
646 BELL SYSTEM TECHNICAL JOURNAL
The relation between the applied electric field, E, and the polariza-
tion induced by it per unit volume is given by
P = i^ E (2)
47r ^ ^
for isotropic materials. The constant (e — l)/47r is the susceptibility
of the dielectric in e.s.u., and e is the dielectric constant, which is
defined as C/Co, where C is the capacitance of the measuring condenser
while it contains the dielectric and Co is its capacitance when empty.
For some purposes there are advantages in considering the actual
polarization, which is produced by a discontinuous distribution of
charged particles, to be replaced by a vector point function which gives
equivalent external effects. Then a vector P may be considered to be
associated with every point in the space occupied by the dielectric and
the dielectric may be considered to have a continuous volume density
of polarization,^ P. In non-isotropic bodies the polarization vector P
induced by an applied field E is not always in the same direction as E,
but is assumed to be a linear vector function * of E (involving, in the
general case, six independent constants), where both E and P are
vector point functions.
In deriving the relationship between the dielectric constant and the
molecular structure of a material it cannot be assumed in general that
the local field which is impressed upon the elementary particles in the
dielectric is simply the field E which can be computed by dividing the
applied voltage V by the distance between the plates of the condenser,
the intensity of the field being assumed to be uniform. For there is an
interaction between the molecules of the dielectric such that each mole-
cule exerts a force on every other molecule. In the absence of an
applied electric field these forces combine with other influences to
create a distribution for which the polarization per unit volume has
the value Po (frequently zero, as has been mentioned). Then when
a field is applied each element of volume in the dielectric is put into a
polarized condition and in general the forces which it exerts upon the
particles in other volume elements changes, because the charges in
each volume element have been displaced to new positions. Conse-
quently, the value assumed by P in a given cube of Fig. 2 will depend
not only upon the direct action of the charges on the plates of the
condenser — which determines the strength of the field E — but also
* Cf. Mason and Weaver, loc. cit. Chap. III.
* Cf. P. Debye, "Polar Molecules," Chemical Catalogue Co., New York (1929),
pp. 32-35.
DIELECTRIC PROPERTIES OF INSULATING MATERIALS 647
upon their indirect action through the polarization which they create
in other elements of volume.
The contribution which the polarization of the dielectric makes to
the force upon a charged particle in it has been calculated by Lorentz
to be (47r/3)P, where P is the polarization per unit volume induced by
the applied field. This calculation applies to an array of particles
with cubic symmetry and to isotropic materials.^ The internal or local
field F is then given by
F = E + ^P. (3)
E may be thought of as the force which has its origin in the direct
interaction between the charges on the plates of the condenser and the
charges in the polarizable complex on which attention has been fixed
(such as one of the cubes of Fig. 2), while the term (47r/3)P may be
regarded as an indirect force coming from the other parts of the dielec-
tric by virtue of their polarized state.
It is assumed in the theory of dielectrics that the structure of mate-
rials is such that P is a linear function of F (or a linear vector function
in the case of anisotropic materials) ; then
P = kF, (4)
where k is the polarizability per unit volume. It can be seen that
E
1 - Ak
(4a)
where A = Air 13, and consequently that the relation between the
polarizability k and the susceptibility (e — l)/4x (= K) is
whenever (3) is a valid expression for the internal field.
The susceptibility can be calculated without presupposing the
validity of equation (3) for the internal field, while the value of k depends
upon whether (i) or some other expression gives the strength of the internal
field in the dielectric.
If L is the number of molecules per cubic centimeter, kjL{= a) is
the polarizability per molecule. This molecular constant a is called
the polarizability of the molecule. By multiplying a. by Avogadro's
number iV, we obtain the polarizability per mole of the dielectric:
6H. A. Lorentz, "The Theory of Electrons," p. 138, and Notes 54 and 55.
648 BELL SYSTEM TECHNICAL JOURNAL
Na = Nk/L. And if m is the mass of a molecule, Nm = M, where M
is the molecular weight, and Lm = p, where p is the density; so that
L p
and the polarizability per mol may be written as Mkjp.
From equations (3) and (4) (or 46) it can be shown that the polariz-
ability is related to the dielectric constant by the familiar relation
* = .-
47r
^A
(5)
which however is only valid when (3) is valid — and for some materials
(3) is apparently not valid.
For gases the term (47r/3)P in (3) is so small as compared with E
that F is approximately equal to E and
i=ir = 4^. (6)
The polarizability and susceptibility are then equal. The physical
reason for this is that the ratio of intermolecular space to the space
occupied by molecules is much larger in a gas than in a solid or liquid
and the direct force exerted by the charges on the condenser plates on
a charged particle in the dielectric is then much greater than the in-
direct force which they exert through the polarization induced in other
molecules.
It is customary to call the quantity (47r/3)^ the volume polarization,
and it is often denoted by the letter p. The volume polarization may
be thought of as 4r/3 times the polarization induced in the dielectric
per unit volume per unit applied field. The convenience of using
(4irl3)k instead of k comes from the occurrence of the factor 47r/3 in
the relation (5) between dielectric constant and polarizability.
On dividing equation (5) by the density we obtain a quantity which
is called the mass polarization, as it is 47r/3 times the polarizability
per gram :
1
li
+ 2
*i-- (7)
6 p
And on multiplying (7) by the molecular weight of the material we
obtain
M
p
- 1
47r M , 47r kN 47r,, ,„,
DIELECTRIC PROPERTIES OF INSULATING MATERIALS 649
The quantity {4:T/3)Na is the molar polarization, Na being the polariz-
abiUty per mole.^
Equation (8), and also (7), expresses the Clausius-Mosotti relation
when a is considered to be a constant characteristic of the individual
molecule and independent of density. The function of e on the left-
hand side of (8) is independent of density whenever a is independent of
density.
The following relation, analogous to that of Clausius and Mosotti
but expressed in terms of the refractive index n, was derived by Lorentz
and by Lorenz :
p n^ -\- 2 S
The left-hand member of this equation is called the molar refraction.
Equations (8) and (8a) are equivalent because of the general relation
between refractive index and dielectric constant (n^ = e), but owing
to the fact that refractive indices are measured at optical frequencies
the molar refraction contains only the electronic part of the total molar
polarization of the material. Subtracting the molar refraction from
the total molar polarization, is one of the methods of determining the
amount of polarization contributed by non-electronic polarizations.
It has been found that the Clausius-Mosotti relation is not equally
satisfactory for all kinds of dielectric polarization. It gives good
results when applied to electronic and atomic polarizations. For
example, in an interesting paper on materials of high dielectric con-
stant, Frank ^ has recently shown that the Clausius-Mosotti-Lorentz-
Lorenz relationship aids materially in explaining the behavior of the
dielectric constants of crystalline materials of high dielectric constant
where the dielectric constant depends upon electronic polarizations.
Where the polarizability of a molecule is the sum of the polarizabilities
of the atoms of which it is composed it is to be expected that if the
relation (5), or (8) or (8a) is valid the sum of the atomic polarizations
would be equal to the molar polarization. Experimental agreement
^ The polarizabilities of non-polar molecules and atoms are usually of the order
of magnitude of lO"^'* c.c, and the molar polarizations of such substances, conse-
quently, are of the order of magnitude of a few c.c, since the molar polarization is
(47r/3) X 6.06 X 10^3 times the polarizability of the individual molecule. The
polarizability of a conducting sphere is equal to the cube of its radius. And, as
atomic dimensions are of the order of magnitude of 10~* cm., it is evident that the
polarizabilities of atoms tend to be of a similar order of magnitude to the polarizabil-
ities which would be expected if they behaved as conducting spheres, though there
are large differences in the ratio of polarizability to volume for different atoms. The
molar polarizations of polar molecules are in general larger than those of similar non-
polar molecules and may be a few hundred c.c. (Cf. P. Debye, "Polar Molecules,"
pp. 12-19.)
7 F. C. Frank, Trans. Faraday Society, 23, (4), 513 (1937).
650 BELL SYSTEM TECHNICAL JOURNAL
with this requirement has been found in optics where the refractive
indices of molecules can be calculated approximately from the molar
refraction (eq. 8a) obtained by adding the atomic refractions.
This additive property of electronic polarizations has been employed
by Frank ^ to interpret the tendency of crystalline materials hav-
ing high dielectric constants to be characterized by a high polariz-
ability/volume ratio for the atoms or ions of which they are composed.
This condition would tend to allow the largest number of highly
polarizable particles to be concentrated in a given space, giving, on
the additivity rule, a high molar polarization and a high dielectric
constant.
On the other hand Wyman ^ has pointed out that the Clausius-
Mosotti relation is not satisfactory when applied to highly polar liquids,
such as water, and has found that for these substances it appears to
be more satisfactory to consider that the polarization is related to the
dielectric constant by the empirical relation
^+^ ^""k. (9)
8.5 3
The calculation of the internal field by Lorentz, which provides the
theoretical basis for equation (8), was made before the theory of polar
molecules had been developed, but equation (8) has since been applied
tentatively to polar molecules."* The problem of obtaining an im-
proved relationship between polarizability and dielectric constant for
materials having molecules with permanent electric moments has been
studied in recent years by several investigators. ^^ The calculation of
the internal field usually involves the assumption that the efifect
of the molecules included in a small sphere surrounding the central
molecule on which the force is being calculated is negligible on the
average because of the random motions due to thermal agitation. On
the supposition that such an assumption is not justified in a polar
material because of the interactions of adjacent polar molecules,
Onsager ^^ has obtained a relation between polarizability and dielectric
constant which for high dielectric constants is nearly the same as
Wyman's empirical relation, equation (9). A comprehen.sive study of
the efifects of interaction between the dipoles of polar molecules has
8 Lqp Clt
^Cf. Wyman, Jour. Amcr. Chein. Soc, 56, 539 (1934); 58, 1482 (1936).
i» Cf. Debye, loc. cit., p. 13.
" Cf. Onsager, Jour. Anier. Chem. Soc, 58, 1486 (1936); Van Arkel and Snoek,
Trans. Faraday Soc, 30, 707 (1934); Wyman, Jour. Amer. Chem. Soc, 58, 1482
(1936); Van Vleck, Jour. Chem. Physics, 5, 320 (1937) and 5, 556 (1937).
'^ Loc. cit.
DIELECTRIC PROPERTIES OF INSULATING MATERIALS 651
been made by Van Vleck by the methods of statistical mechanics. He
obtains an expression which agrees to a second approximation with
that obtained by Onsager. Thus it seems that for highly polar liquids
the relations between polarization and dielectric constant developed
by Onsager, Wyman and Van Vleck may be more satisfactory than
the Clausius-Mosotti relationship, though for many other materials
the Clausius-Mosotti relationship is apparently valid or approximately
valid.
In deriving expressions for the dependence of dielectric constant on
frequency later in this article the formulae obtained will naturally
depend upon which of the equations, (5), (6) or (9), is taken as the
relationship between polarizability and dielectric constant. The
alternative expressions will be listed.
Derivation of a Dispersion Formula
The above-described relations between polarization and dielectric
constant provide the means of obtaining expressions for the variation
of dielectric constant with frequency when we have determined the
dependence of polarizability on frequency. As our object is to exhibit
the general features of anomalous dispersion shared by several par-
ticular types of polarization, it will be sufificient to derive dispersion
formulae containing constants the values of which are not specified,
but which have a sufficiently obvious physical significance. The
derivation given will parallel that of Lorentz in deriving a formula for
optical dispersion, ^^ and in fact is simply a special case of it in which
certain terms are considered to be negligible by comparison with others.
An analogous procedure was used in one of the earliest attempts to
explain anomalous dispersion in the electrical frequency range, the
theory proposed by Drude '* in 1898. This theory was based upon the
hypothesis that anomalous dispersion in the electrical frequency range
depends upon a mechanism similar to that to which optical dispersion
was attributed, the difference being that the particles which produce
anomalous dispersion in the electrical frequency range are so large that
some of the terms in the optical dispersion formula can be neglected.
The formula which Drude derived for electrical anomalous dispersion
yield the same form of variation of dielectric constant with frequency
as do the generally accepted theories of the present time, such as the
Debye theory; the differences lie in the expressions given for the con-
"H. A. Lorentz, "The Theory of Electrons," Chapter IV. See also Korff and
Breit, Reviews of Modern Physics, 4, 471 (1932), where a review of the classical theory
of optical dispersion is given.
" P. Drude, Ann. d. Physik, 64, 131 (1898), " Zur Theorie der anomalien elek-
trischen Dispersion."
652 BELL SYSTEM TECHNICAL JOURNAL
stants in the formulae in terms of properties of the material. Another
adaptation of optical dispersion theory to the explanation of dispersion
in the electrical frequency range was proposed by Decombe ^^ in 1912.
He employed the Lorentz electron theory for the dispersion of light as a
basis for the consideration that if the environment of some of the
electrons in dielectrics is suitable their motions in an applied field could
produce anomalous dispersion and dielectric loss in the electric fre-
quency range. A similar simple and arbitrary assumption regarding
the structure of dielectrics will also be employed here. However, it is
not proposed as a theory of dielectric behavior but merely employed as
a comparatively simple means of deriving and discussing relationships
which can be demonstrated as well on a simple and arbitrary model
as on the more complex ones which correspond more closely to the
actual structure of dielectrics. The relation of the constants in the
dispersion formulae which will be derived here to the actual structure
of dielectrics will only be indicated in a general qualitative way for the
purpose of illustrating the physical nature of the processes involved;
no attempt will be made to provide expressions for the dispersion
constants in terms of other observable properties of the material.
In Fig. 2, let the applied potential be V, where V may vary in general
in any way with the time, though in the present discussion it will be
considered to vary sinusoidally with the time; the impressed field
strength is then given by E — V/d. As in the more general discussion
which preceded this, it will be assumed that the imaginary cells
pictured in Fig. 2 contain large numbers of polarizable complexes
consisting of positive and negative charges in equal numbers held
in position by constitutive forces^ — the origin of which need not be
specified for our present purposes — such that if they are displaced a
distance 5 from their initial positions they will experience a force fs,
where /is a constant, tending to restore them to their initial positions;
and that while these charges are in motion as a result of the action of
the impressed field they experience a frictional force rs, where r is a
constant and s is the velocity in the direction of the impressed field:
and, finally that their motion is also retarded by an inertia reaction
ms, proportional to the mass m and the acceleration s of the particles.
The equation of motion for any typical charge e in a polarizable
complex having the above-described specifications is
ms -f rs + /s = eF, (10)
where F is given by equation (3) in materials to which the Lorentz
calculation of the internal field applies, by F = E in the case of gases
1^ L. Decombe, Journal de Physique, (5), 3, 315 (1912).
DIELECTRIC PROPERTIES OF INSULATING MATERIALS 653
and by other expressions — which in some cases may approximate
either to F = E or to F = E + (47r/3)P — for still other materials.
The quantities F and 5, are vectors, but for isotropic materials 5 is in
the same direction as F.
If, following the method employed by Lorentz, we write an equation
of the form (10) for each charged particle in a physically small volume
8 (such as the cubes of Fig. 2), multiply each equation by e, add the
equations for all of the particles in 8, and divide by the volume 8,
we obtain
mP -{- rP +fP = ne^F, (11)
where P = (l/5)X!e5 and n is the number of charged particles charac-
terized by the constants m, r and / per unit volume. The volume 8
may be considered to be that of one of the cubes in Fig. 2. As indi-
cated earlier it should contain a sufftcient number of molecules to give
a good mean value for P, the polarization per unit volume, but at the
same time it should be small enough not to mask significant spatial
variations in P.
\\'hen the impressed field E is varying sinusoidally with the time at
the frequency aj/27r, the local or internal field F tending to displace
each charged particle in the dielectric will also vary sinusoidally with
the time, though in general out of phase with E, if F is given by equation
(3), and can be considered to be given by the real part of Foe'"'. Under
these conditions
P = kFoe"^'
solution of equation (10) for the steady state provided that
k^j-. ""'' ,.,,■ (12)
k is the polarizability per unit volume and is a complex quantity, since
the term zVco in the denominator is an imaginary {i = V — 1).
Equations (10), (11) and (12) apply to a dielectric having a single
type of polarization characterized by the constants /, r, m, n and e.
But in general an applied field induces several types of polarization
simultaneously in a dielectric, and if we assume that it induces w
types which are independent of each other, the total polarization per
unit volume is given by
P = kiF + k.F -\- • • • KF. (13)
The total polarizability is then the sum of the individual polarizabili-
ties, or
w
k = Z k,. (14)
3=- 1
is a
654 BELL SYSTEM TECHNICAL JOURNAL
In this discussion it will be sufficient to consider that the different types
of polarization designated by ki, k^ - • • kw differ from one another only
in having different sets of values for the constants of equation (12),
designated by the subscripts 1,2,3 • • • w; for example, the character
of the polarizability k-i is specified by the set of constants mi, ri,/i and «i.
In the first place it is evident that when the frequency of alternation
of the voltage applied to the dielectric lies in the radio and power range
it is possible to select any number of sets of values of m, r,/ which will
make the terms ww^ and rw negligible in comparison with / in the
denominator of (12). Let mi, ri, /i be an example of such a set of
constants and let there be «i particles per unit volume to which these
constants apply. Then for this type of polarization equation (12)
reduces to
^i-¥- (15)
This type of polarization is independent of frequency and will be
referred to as an instantaneous polarization or an optical polarization.
The main representatives of the instantaneous or optical polarizations
are the electronic and atomic polarizations, which experience dis-
persion in the visible and infra-red but which are independent of fre-
quency in the electrical range, and the contribution of this polariz-
ability to the dielectric constant is therefore frequently calculated from
refractive index measurements.
A second type of polarization results if we assume that the dielectric
we are considering contains a class of particles for which mo:'^ in equation
(12) is negligible by comparison with rw and with/, but in which rw is
of the same order of magnitude as/ in the electrical range of frequencies.
Let W2, ^2, /2 be a typical member of this class, the number of such
particles per unit volume of the dielectric being n-i. Then for this class
of particles equation (12) becomes
{tr2w + /?)
This expression represents the type of variation with frequency to
which the name anomalous dispersion is given, and in the preceding
paper the type of polarization which produces it was called an absorp-
tive polarization.
It can readily be seen also that neglecting the ni's term in (10) or
the mP term in (11) leads to the same expression for k, i.e., equation
(16), as does neglecting the mw^ term in the denominator of (12).
So for any member, (j^, ji, n^, of the class of particles which produces
DIELECTRIC PROPERTIES OF INSULATING MATERIALS 655
anomalous dispersion, equation (10) reduces to
r,s-^f,s = eF (17)
and equation (11) becomes
r2p +hP = n,e'F. (18)
Decombe's theory, which has been mentioned earlier, was based upon
an equation equivalent in most respects to (18), while Drude's expres-
sions for dispersion were obtained by a method equivalent to neglecting
wco^ in (12).
Each term in equations (17) and (18) has an evident dynamical
significance. Consequently, a physical picture of the essential nature
of the anomalous dispersion process is given by equations (17) and
(18) even though the values of constants r2,/2, «2 and e are not specified
in terms of independently measurable properties of the dielectric.
Thus the term f2S represents a restoring force tending to return the
particles displaced by the impressed field to their initial positions, the
constant /2 acting as a stiffness coefficient; the term r2S acts as a fric-
tional force, r being a measure of the friction experienced by, for exam-
ple, a moving ion or a rotating polar molecule; and, finally, eF is the
driving force tending to displace a particle of charge e. Evidently
conditions which are sufficient to produce anomalous dispersion exist
whenever the motion of charged particles in an applied field is suffi-
ciently specified by considering the effects of a restoring force pro-
portional to the displacement of the typical particle and of a frictional
force proportional to the velocity of the particle in the direction of
applied field, as in equation (17). Or, putting it in more general terms,
we may say that anomalous dispersion occurs whenever the relation
between the polarization per unit volume and the force due to the
internal electric field is given by an equation which can be reduced to
(18). However, the possibility that anomalous dispersion may also
occur under conditions which cannot be described by equation (18)
is not excluded by the considerations given here.
A third type of polarization which can be obtained by selecting
suitable sets of values for the constants of equation (12) is that in
which none of the terms in the denominator of (12) can be neglected in
the electrical range of frequencies. Let ks be the polarizability for
this type of polarization which can then be represented by affixing the
subscript 3 to the constants m, r, f and n of equation (12). This type
of dispersion includes both the normal and the anomalous types but,
as has already been indicated, in the radio and power ranges of fre-
656 BELL SYSTEM TECHNICAL JOURNAL
quency examples of a dispersion of this kind have not as yet been
observed in dielectrics which are not piezo-electric.^® It follows then
that dielectrics behave as though the inertia of the particles which
contribute to dielectric polarization is small enough that the inertia
reaction mor can be neglected in the electrical frequency range. This
is an empirical result; the possibility of a polarization of the type kz
occurring in the electrical frequency range is not excluded by the
general theory of dispersion. The higher the frequency of an im-
pressed field the greater should be the likelihood of encountering the
type of frequency-variation described by kz (or equation (12)), because
the prominence of the moP' term increases with the square of the
frequency.
The preceding discussion shows that we can write equation (14)
in the form
k = k, + ka, (19)
where k is the total polarizability, ki is the sum of the instantaneous
polarizabilities and ka the sum of the absorptive polarizabilities, that is,
of the polarizabilities which vary with frequency according to equation
(16). If for simplicity we take the case in which the dielectric has only
one representative of ki and one of ka, we obtain by substituting the
values of ki and ka given respectively in (15) and (16),
k = ^ + ,. ""', (20)
/i (^^2W + /a)
as an expression for the total polarizability.
Defining r' by t' = r/f, and dropping the subscripts in (20) to make
the notation simpler, we obtain
K Ki ~\ p
1
1 + iu^r'
(21)
which is the total polarizability per unit volume for a dielectric having
two types of polarization, the one represented in (21) by the instan-
taneous polarizability ki and the other by the absorptive polarizability
1'^ Piezo-electric crystals such as quartz and Rochelle salt form exceptions, but for
them dielectric polarization is coupled to macroscopic mechanical strains in the
material and the mass reactance is due to the flexing or extension of the entire crystal.
The dielectric constant of such a crystal as measured in almost any direction,
shows an increase with increasing frequency, followed by anomalous dispersion. This
is the behavior required by equation (12), or rather by an equation for the dielectric
constant derivable from equation (12). This dispersion, however, depends upon the
size and shape of the crystal, the nature of the electrodes and the manner of supporting
the crystal during the measurements, and the exact interpretation of such measure-
ments is a rather complex procedure. See, for example, W. P. Mason, Proc. L R. E.,
23, 1252-1263 (1935).
DIELECTRIC PROPERTIES OF INSULATING MATERIALS 657
specified by the second term on the right. The quantity t' is called
the relaxation-time.
On multiplying the left-hand side of equation (21) by (47r/3)(M/p)
and the right-hand side by (4x/3)(7V/L) we obtain
Air Mk _ ^ { ki
.,, ne^ { 1
fL\\^ ic
(22)
which is the molar polarization.
For dielectrics to which the Clausius-Mosotti relation applies,
equation (8) shows that
^ir Mk M e- \
3 p p € -\- 2
(22a)
and in fact the expression on the right-hand side of (22a) is frequently
called the molar polarization. Reference to equation (6) shows, how-
ever, that for gases (22a) reduces to the simpler relation.
4^Mk M(e- 1)
TV^7~1 — ^^^^^
And for Wyman's relation between dielectric constant and polariz-
ability, which has been discussed earlier, the molar polarization be-
comes
4TAIk M e -f 1
3 p p 8.5
(22c)
Equations (22a), (22b) and (22c) are not the only relations between
dielectric constant and molar polarization which have been proposed,
but they apparently cover moderately well many of the conditions
met in practice. For the right-hand member of equation (22) can be
substituted whichever of the three expressions (22a), (22b), (22c) seems
the most suitable for the type of dielectric under investigation.
If in equation (21) w is set equal to zero we obtain the zero-frequency
(or static) polarizability
k^ = ki + ne'^lf (23)
and if w is set equal to infinity we obtain
^oo = ki. (24)
Subtraction gives
ko — k^ — ne^lf. (25)
Substituting (24) and (25) in (21) gives
ko -
TTi
k = k^-^( ^\ .^'^, ) • (26)
658
BELL SYSTEM TECHNICAL JOURNAL
The constants ne^ff and ki are not present in (26), being replaced by
two special values of the polarizability, the zero-frequency value and
the infinite-frequency value. However, it is not the polarizability
but the dielectric constant which is directly observed in measurements
on dielectrics, so it is desirable to replace ^o and ka, by their equivalents
in terms of the dielectric constant. But, as the earlier discussion has
indicated, the relation between dielectric constant and polarizability is
different for different types of dielectrics; three alternative expressions
analogous to (22a), {22b) and (22c) will therefore be derived.
For materials to which equation (22a) (or the equivalent and simpler
relation (5)) applies
^0 — ^oo =
60- 1
eo + 2
r
+ 2.
9(
eo
Coo)
47r(eo + 2){e^ + 2)
(27)
where eo is the zero-frequency dielectric constant and eco is the infinite-
frequency dielectric constant. Then equation (26) can be replaced by
47r
€0
1
+ 2
+
eo- 1
60 + 2
-f 2 J 1 + icor'
(28)
By rationalizing and using the second expression given for ^o — ^od in
equation (27) we can write equation (28) in the alternative form
47r^
+ 2
+
3(60
.)
(60 + 2)(6oo + 2)
— i
1
1 + coV'
3(€o — 6oo)
(eo + 2)(€^ + 2)J l-fcoV'
(29)
Equation (29) is the complex polarizability per unit volume multiplied
by the factor 47r/3 and expressed in terms of observable values of the
dielectric constant and the relaxation-time t'. The relaxation-time
can also be expressed in terms of the reciprocal of a special value of the
frequency; this permits all of the theoretical constants such as ne^/f
and t' to be replaced by certain special values of the dielectric constant
and a critical value of the frequency,
A simpler expression for the polarizability is obtained in the case of
gases, or whenever equation (6) gives the relation between polariz-
ability and dielectric constant. Equation (26) then gives
4.Trk 1
- 1 +
60 — 6o
(30)
1 + iur'
And for materials to which the relation (cf. equation (9)) proposed by
DIELECTRIC PROPERTIES OF INSULATING MATERIALS 659
Wyman applies the procedure followed above yields
On multiplying equations (29), (30) and (31) by Mjp three alterna-
tive formulae for the molar polarization of a dielectric having polariza-
tions of the type specified by equation (21) are obtained ; the constants
in these formulae include only special values (eo and fa,) of the dielectric
constant and the relaxation-time, all of which can be obtained from
dispersion curves.
The quantity ko — ^00 is a constant of the material, which, as equa-
tion (26) shows, represents the largest value which the absorptive part
of the total polarizability, i.e., the ka term in (19), can have for a given
material ; it may be described as the zero-frequency or static value of
the absorptive part of the polarizability. Evidence as to the nature
of a polarization can be obtained by investigating experimentally the
dependence of (^0 — k^)lp on temperature; for example, if the polariza-
tion is due to the changes of orientation of polar molecules according
to the Debye theory this quantity should increase linearly with the
reciprocal of the absolute temperature. It is useful, therefore, to
express (^0 — ^co)/p in terms of observable values of the dielectric
constant so that it may be plotted against temperature. In this con-
nection there is, however, the same complication which has appeared
in other places in this discussion regarding the relation between dielec-
tric constant and polarizability. The three relations which have been
discussed here yield for (^0 — ^co)/p the following expressions:
3 r eo - 1 600-1
(^0 - U/P =
47rp [ eo + 2 ecx, + 2
(Clausius-Mosotti)
(32a)
(for gases)
(32^)
(Wyman).
(32c)
47rp \ 3
— 3 / eo — €0
~ 47rp \ 8.5
The Complex Dielectric Constant
As the dielectric constant (e) is the quantity directly measured in
experimental investigations it is desirable to determine how it should
vary with frequency for the type of dielectric polarization described in
equation (21) or (29). Solving equation (5) for e we obtain
1 +8^^
e = —. i^i)
l-4f.
660
BELL SYSTEM TECHNICAL JOURNAL
By substituting the expression for 4(7r/3)^ given in equation (28) into
(33) we obtain
+ t
eo + 2
+ 2
eo + 2 €oo + 2
(34)
eo
eo + 2 Coo ,
Eo + 2 \ Coo + 2 eo
1
eo + 2
1 + t p^ COT
Coo + 2
Then, by setting
we obtain
+ 2 ,
+ 2^
= T,
e
=
(l +i
— cor
eo /
eo
1 +
iiiiT
(34a)
(35)
(36)
and transforming this into polar form to faciHtate division gives
£ _ Plg"^l ^ pl
eo P2e'*'2 P2
where
Pl =
cos (v?! — v'2) + ^sin (v?! — (^2)
1 + I I w-r-
eo
P2 = [1 + coV2]s
(37)
^Pl = tan~^ — cor and ) and between t' and t
(fco - feco) t'
^, • 1>T . , • 13 , .• 3 3(€0 — €co) €„, "t" 2
Llausius-Mosotti Relation — - • -. , ^. . ; — — - ■ ■ — r
47r (€0 -h 2)(ecD -|- 2) to + 2
Gases —- • (to — «co) t
47r
■2
Wyman's Empirical Relation -— • — "" t
47r 8.5
and T in Table I. The resulting expressions can then be substituted
for (^0 — ^co) and r' in equation (26) yielding expressions for the polariz-
abilities of the different types of polarization listed in Table I. The
molar polarization can then be obtained by multiplying (26) by
(47r/3)(M/p). However, in general it is not likely that any useful
purpose is to be served by calculating the molar, polarization for inter-
facial polarizations; a more significant quantity would be the polariza-
tion per conducting particle, when the polarization is of the type {M),
Table I, and the number of conducting particles per unit volume can
be estimated.
We have pointed out that a number of theories which have been pro-
posed for the explanation of the variation of dielectric constant with
frequency may be expressed in the forms (41a) and (416) when the
expressions listed in Table I are substituted for (eo — eco) and r, but
we have not yet indicated how these formulae agree with experimental
data. For such materials as ice (see Fig. 3, for example) and for
666
BELL SYSTEM TECHNICAL JOURNAL
certain alcohols and glycols, the experimental points agree fairly
closely with the curves obtained by plotting equations (41a) and (416)
for a suitable choice of the values of the constants.
But for many other dielectrics, particularly non-homogeneous
systems or disperse systems such as those listed under Item 3, Table I,
the simple dispersion formulae (41a) and (416) often fall very far short
of adequately representing the experimental data. Von Schweidler ^o
and Wagner ^^ have attempted to explain the form of dispersion curves
obtained for such materials by postulating that the polarizations
^ 70
=v
^,
\
s
\
e"
\
y
\\
\,
\
\
\
\
\
\
\
-12.0'
^
\
*-
-7.1 = DEGREES
' 1 CENT.
^
\
/>
<
\
s
>
\
\
\
\
\
•
•
•
t
/
/
\
\
>
\ \
'\
/
\
V
X
\cS>
\
y
\
V
\\
\ V
V \
\
^v
\
\
>
N
s.
V
^ \^
-"^
•V
N.
~
>4.3
>.
^
"^^^
^^
;^
^
^^ ^V.
1
—
,~
- — — —T
■■»-~
^.
_
50 100 500 1000 5000 10,000
FREQUENCY IN CYCLES PER SECOND
Fig. 3 — Experimental dispersion curves for ice.
100,000
induced in the dielectric have a wide range of relaxation times at any
given temperature, instead of a single relaxation time, as for the
polarizations listed in Table I. A further contribution to the theory
of the distribution of relaxation times has recently been made by
Yager .^^ However, in spite of the existence of many materials which
do not show the type of dispersion described by (41a) and (416), the
value of these formulae in interpreting experimental data is consider-
able, particularly as applied to pure materials.
Table I emphasizes the point that mere agreement of experimental
data for dielectric constant and dielectric loss with the theoretical
20 E. V. Schweidler, Aym. d. Phys., (4) (24), 711 (1907).
21 K. W. Wagner, Archivf. Elektrotechnik, 2, 371 (1914j.
22 W. A. Yager, Physics, 7, 434 (1936).
DIELECTRIC PROPERTIES OF INSULATING MATERIALS 667
curves obtained by plotting (41a) and (416) for suitably adjusted
values of the constants only places the type of mechanism to which
the observed dispersion can be attributed within the rather large
catagory which includes at least the seven types of mechanism listed in
the table. Data showing the dependence of (^o — ^co)/p on temperature
allows a further specialization of the processes which could account
for the observed behavior; and of course a number of possibilities can
be discarded on general grounds of physical improbability . And
finally, agreement of the constants calculated from dielectric measure-
ments with the values calculated from independent estimates of the
sizes and other characteristics of the molecules or other elementary
units which contribute to the polarization provides the most convincing
evidence of the nature of the polarization. Such agreement is fre-
quently obtained in the application of the Debye theory to gases and
liquids.
The characteristics which can be deduced from equations (41a) and
(416) without substituting for the constants theoretical expressions,
such as those given in Table I, are of considerable value in interpreting
electrical measurements upon dielectrics. It may be convenient to
describe these as the general characteristics of anomalous dispersion,
distinguishing them thereby from the special characteristics peculiar
to particular kinds of dielectric polarization which share the property
of producing anomalous dispersion in the radio and power range of
frequencies.
Appendix
The following list contains the definitions of the quantities which
appear in Table I :
ei, €2, 7i, 72 are respectively the dielectric constants and conductivities
of two materials designated by subscripts 1 and 2, the unit
of conductivity being such that 7 = 367r X 10'^ X, where
X is in (ohm-cm)~^
eo, Coo are respectively the dielectric constant at the lower and
upper extremities of dispersion curves; they are called the
zero-frequency (or static) dielectric constant and the
infinite-frequency dielectric constant,
L is the number of molecules per unit volume,
y] the viscosity of a liquid containing polar molecules,
k Boltzmann's constant,
T the absolute temperature,
IX the permanent electric moment of a polar molecule,
668 BELL SYSTEM TECHNICAL JOURNAL
a the radius of a polar molecule, assumed to be spherical,
h the radius of a colloidal particle,
d the thickness of a conducting skin on the particle of radius h,
r a frictional resistance coefficient of unspecified origin,
/ an elastic restoring force coefficient of unspecified origin,
n the number per unit volume of elementary charged particles
subject to certain specified conditions,
p the ratio of the volume occupied by the spherical particles
in (3c, d, e) to the total volume,
C\ the capacity of the blocking layer of (36), Table I,
R the resistance of the dielectric, exclusive of the blocking
layer.
The following list contains the definitions of quantities which appear
in other parts of the article.
CO is 1-K times the frequency of alternation of the applied field,
V the applied voltage,
R the intensity of the applied field,
P the polarization per unit volume induced by a field £,
F the internal or local field,
p the density of the dielectric,
M the molecular weight of the material of which the dielectric is
composed,
m the mass of a molecule; in another context, the mass of any charged
particle considered in the discussion,
N is Avogadro's number, 6.06 X 10'^ molecules per mole,
5 the displacement of a charged particle from an equilibrium position
by an applied field,
k the velocity of the charged particle in the applied field,
s the acceleration of the particle in the applied field.
References Relating to Table I
1. P. Debye, "Polar Molecules," New York (1929).
2. P. Drude, Ann. d. Physik, 64, 131 (1898); L. Decombe, J. d. Physique (5) J, 215
(1912); and the present article.
3. K. W. Wagner, Chap. I of Schering's "Die Isolierstoffe der Elektrotechnik,"
Springer, Berlin (1934).
4. A. Joffe, "The Physics of Crystals," New York (1928).
5. K. W. Wagner, Arch. f. Elektrotechnik, 2, 371 (1914).
6. J. B. Miles and H. P.'Robertson, Phys. Rev., 40, 583 (1932).
7. A. Geniant, "Die Elektrophysik der Isolierstoffe," Berlin (1930).
General reviews of the theory of dielectric behavior as it concerns dispersion for
power and radio frequencies are included in the following places, among others:
1. E. Schrodinger, " Dielektrizitat," Graetz, Handb. d. Elek. u. d. Magn., Leipzig
(1918), pp. 157-229.
DIELECTRIC PROPERTIES OF INSULATING MATERIALS 669
2. E. V. Schweidler, " Die Anomalien der dielektrischen Erscheinungen," ibid., p. 232 ;
Ann. d. Phys. (4) 24, 711 (1907).
3. J. B. Whitehead, "Lectures on Dielectric Theory and Insulation," McGraw-Hill
(1927).
4. L. Hartshorn, Jour. I. E. E., 64, 1152 (1926).
5. P. Debye, "Polar Molecules," Chem. Cat. Co., New York (1929).
6. Schering's "Die Isolierstoffe der Elektrotechnik," Springer, Berlin (1924).
7. A. Gemant, "Die Elektroph)sik der Isolierstoffe," Berlin (1930).
Abstracts of Technical Articles from Bell System Sources
Electron Microscope Studies of Thoriated Tungsten} Arthur J.
Ahearn and Joseph A. Becker. Many past experiments have shown
that the thermionic activity of a thoriated tungsten filament is deter-
mined by the concentration of thorium on its surface. This concentra-
tion is in turn determined by the rate of arrival and rate of evaporation
of thorium. Typical published values of these rates are given in Fig. 1.
An electron microscope used to obtain electron images of thoriated
tungsten ribbons is described. Comparison with photomicrographs
shows that the active and inactive patches composing an electron image
agree in size, shape and number with the exposed grains of the tungsten.
The electron microscope shows that thorimn comes to the surface in
''eruptions'' at a relatively small number of randomly located points.
From a comparison of photomicrographs showing thoria globules and
electron images of thorium eruptions, it is deduced that all the thorium
in a globule comes to the surface when an eruption occurs. Factors such
as a high temperature flash and sudden heating and cooling of the
filament affect the frequency of eruptions. Thorium eruptions are the
only observed manner in which thorium arrives at the filament surface.
They are repeatedly observed in the early stages of thoriation. Erup-
tions are not observed in the later stages of thoriation where con-
ditions are unfavorable for their observance. Electron images of a
Pintsch single crystal filament reveal alternate active and inactive
bands parallel to the filament axis. Thorium eruptions occur only on
the active bands. With a polycrystalline ribbon the surface migration
of thorium from the eruption centers is isotropic; with a single crystal
ribbon there is a strongly preferred direction of migration. X-ray
analysis shows that the surface is a (211) plane and that the preferred
direction of migration agrees with the (111) direction in this plane.
During the process of thoriating a filament the relative emissions from
different grains change by substantial amounts; in many cases the
change is so great that the relative emissions are reversed. Measure-
ments of work function differences between grains gave values ranging
up to 0.6 volt.
The Mechanism of Hearing as Revealed through Experiment on the
Masking Effect of Thermal Noise.^ Harvey Fletcher. In an electri-
1 Phys. Rev., September 15, 1938.
''Proc. Nat'l. Acad. Set., July 1938.
670
ABSTRACTS OF TECHNICAL ARTICLES 671
cal conductor there is a statistical variation of the electrical potential
difference between its two ends which is due to the thermal agitation
of the atoms, including the electrons. This electrical noise is amplified
by means of a vacuum tube amplifier and then converted into an
acoustical noise by means of a telephone receiver held on the ear.
When this noise is present it reduces the capability of the ear to hear
other sounds. The intensity per cycle of the acoustical noise compared
to the intensity of a pure tone which can just be perceived in the pres-
ence of a noise was determined experimentally using a group of ob-
servers. This relative intensity for a given frequency range was
constant throughout a wide variation of intensity. However, its
value does vary with the position in the frequency spectrum and it is
the amount of this variation which enables one to calculate the relation
between the frequency of the tone and its position of maximum stimula-
tion along the basilar membrane. The results of such a calculation
are given and shown to be in good agreement with determinations from
animal experimentation.
Transcontinental Telephone Lines. ^ J- J- Pilliod. A fourth trans-
continental line has just been created by the completion of four pairs
of open wire between Oklahoma City and Whitewater, California.
This open-wire line connects at its eastern terminus with the already
existing toll cables from the east, and at its western terminus with a
toll cable running into Los Angeles.
In a cross-section of the United States just west of Denver, there
are now 140 through telephone circuits and about the same number
of telegraph circuits carried by four open-wire routes.
The four new pairs which constitute the transcontinental line carry,
in addition to the usual voice frequency channels, three channels of
carrier. But their design throughout has been such that twelve
additional carrier circuits can be superimposed upon the four channels
now provided by each wire pair.
The wires of each pair are spaced 8 inches apart with the nearest
spacing between pairs being 26" while crossarms are 36" apart. New
transposition systems have also been used to further reduce crosstalk.
Application of Statistical Methods to Manufacturing Problems."^ W.
A. Shewhart. The application of statistical methods in mass produc-
tion makes possible the most efficient use of raw materials and manu-
facturing processes, effects economies in production, and makes
possible the highest economic standards of quality for the manufac-
tured goods used by all of us. The story of the application, however,
^ Electrical Engineering, October 1938.
* Jour. Franklin Institute, August 1938.
672 BELL SYSTEM TECHNICAL JOURNAL
is of much broader interest. The economic control of quahty of
manufactured goods is perhaps the simplest type of scientific control.
Recent studies in this field throw light on such broad questions as:
How far can Man go in controlling his physical environment? How
does this depend upon the human factor of intelligence and how upon
the element of chance?
Observational Significance of Accuracy and Precision} W. A.
Shewhart. Two of the most common terms used in pure and applied
science are accuracy and precision. When such terms are used, as in
the specification of quality of manufactured products, it is desirable
that they have definite and, in so far as possible, experimentally veri-
fiable meanings. It is, therefore, important to determine how far one
can go towards attaining this end by applying with rigor the principle
that only that which is observable is significant. In the application
of the concepts of accuracy and precision, it is customarily assumed
that the available data constitute a random sample. Hence, the first
step in attaining experimentally verifiable meaning of these terms is to
choose an operationally verifiable criterion of randomness. One such
criterion is the quality control chart. In order to give experimental
definiteness to any measure of either accuracy or precision derived
from a random sample, it is also necessary to specify the way any
statement involving the measure may be experimentally verified. To
do this it is necessary to make at least four empirical choices as to the
details of taking and analyzing the data in the process of verification.
Hence, it appears that the meaning of either precision or accuracy is
verifiable. Hence, it appears that the meaning of either precision or
accuracy is verifiable only in a limited sense subject in any specific case
to the choice of empirical criteria of verification.
The Time Lag in Gas- Filled Photoelectric Cells} A. M. Skellett.
In commercial gas-filled photoelectric cells there is a lag in response
which becomes appreciable above frequencies in the neighborhood of
10,000 cycles. If this lag is due to the transit times of the ions across
the cell, it should be possible to set up resonance conditions by varying
the frequency of modulation of the incident light intensity. This has
been accomplished in a cell of special design and the resonance condi-
tions agree with the theory, thereby demonstrating that the transit
time of the ions is the simple cause. The paper also discusses the flow
of the ions and electrons across the cell and their impacts in relation
to the flow of current in the external circuit.
5 Jour. Wash. Acad. Sciences, August 15, 1938 (p. 381).
^ Internat'l. Projectionist, September 1938; Jour. Applied Physics, October 1938.
Contributors to this Issue
Charles R. Burrows, B.S. in Electrical Engineering, University
of Michigan, 1924; A.M., Columbia University, 1927; E.E., Univer-
sity of Michigan, 1935. Research Assistant, University of Michigan,
1922-23. Western Electric Company, Engineering Department,
1924-25; Bell Telephone Laboratories, Research Department, 1925-.
Mr. Burrows has been associated continuously with radio research and
is now in charge of a group investigating the propagation of ultra-short
waves.
Arthur B. Crawford, B.S. in Electrical Engineering, Ohio State
University, 1928. Member of Technical Staff, Bell Telephone
Laboratories, 1928-. Mr. Crawford has been engaged chiefly in work
relative to radio communication by ultra-short waves.
Carl R. Englund, B.S. in Chemical Engineering, University of
South Dakota, 1909; University of Chicago, 1910-12; Professor of
Physics and Geology, Western Maryland College, 1912-13; Laboratory
Assistant, University of Michigan, 1913-14. Western Electric Com-
pany, 1914-25; Bell Telephone Laboratories, 1925-. As Radio
Research Engineer Mr. Englund is engaged largely in experimental
work in radio communication.
L. A. Meacham, B.S. in Electrical Engineering, University of Wash-
ington, 1929. Cambridge University, England, 1929-30. Bell Tele-
phone Laboratories, 1930-. Mr. Meacham's work has been concerned
with the generation and distribution of constant reference frequencies.
S. O. Morgan, B.S. in Chemistry, Union College, 1922; M.A.,
Princeton University, 1925; Ph.D., 1928. Western Electric Company,
Engineering Department, 1922-24; Bell Telephone Laboratories,
1927-. Dr. Morgan's work has been on the relation between dielectric
properties and chemical composition.
William W. Mumford, B.A., Willamette University, 1930. Bell
Telephone Laboratories, 1930-. Mr. Mumford has been engaged in
radio receiving work, chiefly on the problem of propagation and
measurement in the ultra-short-wave region.
E. J. Murphy, B.S., University of Saskatchewan, Canada, 1918;
McGill University, Montreal, 1919-20; Harvard University, 1922-23.
673
674 BELL SYSTEM TECHNICAL JOURNAL
Western Electric Company, Engineering Department, 1923-25; Bell
Telephone Laboratories, 1925-. Mr. Murphy's work is largely con-
fined to the study of the electrical properties of dielectrics.
A. C. NoRWiNE, A.B., University of Missouri, 1923; B.S. in Elec-
trical Engineering, 1924; E.E., 1925. Bell Telephone Laboratories,
1925-. Mr. Norwine has been principally engaged in studies of the
effects of transmission delay and voice operated devices on toll tele-
phone circuits.
A. J. Rack, B.S. in Electrical Engineering, University of Illinois,
1930; M.A. in Physics, Columbia University, 1935. Bell Telephone
Laboratories, 1930-. Starting with radio research, Mr. Rack has more
recently been engaged in the analysis of special problems arising in
amplifier circuits.
E. F. Watson, M.E., Cornell University, 1914. American Tele-
phone and Telegraph Company, Engineering Department, 1914-19;
Department of Development and Research, 1919-34. Bell Telephone
Laboratories, 1934-. Mr. Watson has been concerned with the de-
velopment of various types of telegraph equipment, particularly
teletypewriters, telephotograph equipment, telegraph maintenance
and testing equipment, grounded telegraph systems and regenerative
telegraph repeaters. His present work as Teletypewriter Engineer is
along these same lines.
S. B. Wright, M.E. in Electrical Engineering, Cornell University,
1919. Engineering Department and Department of Development and
Research, American Telephone and Telegraph Company, 1919-34;
Bell Telephone Laboratories, 1934-. Mr. Wright is engaged in trans-
mission development of radio systems.
Index to Volume XVII
Alniqidst, M. L., H. J. Fisher and R. H. Mills, A New Single Channel Carrier Tele-
phone System, page 162.
Amplitude Characteristics of Telephonic Signals, Devices for Controlling, A. C.
Nor wine, page 539.
Amplitude Range Control, S. B. Wright, page 520.
Analyzer, An Optical Harmonic, H. C. Montgomery, page 406.
Anti-sidetone Subscriber Set, Common Battery, An Explanation of the, C. 0. Gibbon,
page 245.
B
Best, F. H., New Transmission Measuring Systems for Telephone Circuit Main-
tenance, page 1.
Blanchard, Julian, Hertz, the Discoverer of Electric Waves, page 327.
Bode, H. W., Variable Equalizers, page 229.
Burrows, Chas. R., The Exponential Transmission Line, page 555.
Cable Carrier System, Crystal Channel Filters for the, C. E. Lane, page 125.
Cable Carrier Telephone System, Crosstalk and Noise Features of, M. A. Weaver,
R. S. Tucker and P. S. Darnell, page 137.
Cable Carrier Telephone Terminals, R. W. Chesnut, L. M. Ilgenfritz arid A. Kenner,
page 106.
Cable System for Television Transmission, Coaxial, M. E. Slriehy, page 438.
Cables, Toll, A Carrier Telephone System for, C. W. Green and E. I. Green, page 80.
Carr, J. A. and F. V. Haskell, Studies of Telephone Line Wire Spacing Problems,
page 195.
Carrier Telephone System for Toll Cables, A, C. W. Green and E. I. Green, page 80.
Carrier Telephone System, Cable, Crosstalk and Noise Features of, M. A. Weaver,
R. S. Tucker and P. S. Darnell, page 137.
Carrier Telephone System, A New Single Channel, H. J. Fisher, M. L. Almquisi and
R. H. Mills, page 162.
Chesnut, R. W., L. M. Ilgenfritz and A. Kenner, Cable Carrier Telephone Terminals,
page 106.
Clarke, Beverly L. and A. E. Ruehle, Spectrochemical Analysis in Communication
Research, page 381.
Coaxial Cable System for Television Transmission, M. E. Strieby, page 438.
Crawford, A. B., C. R. Engl und and W. W. Mumford, Ultra-Short-Wave Transmission
and Atmospheric Irregularities, page 489.
Crystal Channel Filters for the Cable Carrier System, C. E. Lane, page 125.
Darnell, P. S., M. A. Weaver and R. S. Tucker, Crosstalk and Noise Features of
Cable Carrier Telephone System, page 137.
Darrow, Karl K., Radioactivity — Artificial and Natural, page 292.
Davisson, C. J., The Discovery of Electron Waves, page 475.
Dielectric Properties of Insulating Materials, The, E. J. Murphy and S. 0. Morgan,
page 640.
Diodes, Effect of Space Charge and Transit Time on the Shot Noise in, A. J. Rack,
page 592.
E
Echo Suppressors, Two, The Occurrence and Effect of Lockout Occasioned by,
Arthur W. Horton, Jr., page 258.
5
BELL SYSTEM TECHNICAL JOURNAL
Electric Waves, Hertz, the Discoverer of, Julian Blanchard, page 327.
Electron Waves, The Discovery of, C. J. Davisson, page 475.
Englund, C. R., A.B. Crawford and W. W. Mumford, Ultra-Short- Wave Transmission
and Atmospheric Irregularities, page 489.
Equalizers, Variable, H. W. Bode, page 229.
Fay, C. E., A. L. Samuel and W. Shockley, On the Theory of Space Charge between
Parallel Plane Electrodes, page 49.
Feedback Oscillators, Stabilized, G. H. Stevenson, page 458.
Filters, Crystal Channel, for the Cable Carrier System, C. E. Lane, page 125.
Fisher, H. J., M. L. Almquist and R. H. Mills, A New Single Channel Carrier Tele-
phone System, page 162.
Gibbon, C. O., An Explanation of the Common Battery Anti-sidetone Subscriber Set,
page 245.
Green, C. W. and E. L, A Carrier Telephone System for Toll Cables, page 80.
Gustafson, W. G., Magnetic Shielding of Transformers at Audio Frequencies, page 416.
Haskell, F. V. and J. A. Carr, Studies of Telephone Line Wire Spacing Problems,
page 195.
Herriott, W., High Speed Motion Picture Photography, page 393.
Hertz, the Discoverer of Electric Waves, Julian Blanchard, page 327.
Horton, Arthur W., Jr., The Occurrence and Effect of Lockout Occasioned by Two
Echo Suppressors, page 258.
I
Ilgenfritz, L. M., R. W. Chesnut and A. Kenner, Cable Carrier Telephone Terminals,
page 106.
Impedance Concept and its Application to Problems of Reflection, Refraction,
Shielding and Power Absorption, The, 5. A. Schelkunoff, page l7.
Inglis, A. H., Transmission Features of the New Telephone Sets, page 358.
Insulating Materials, The Dielectric Properties of, E. J. Murphy and S. 0. Morgan,
page 640.
J
Jones, W. C, Instruments for the New Telephone Sets, page 338.
K
Kenner, A., R. W. Chesnut and L. M. Ilgenfritz, Cable Carrier Telephone Terminals,
page 106.
L
Lane, C. E., Crystal Channel Filters for the Cable Carrier System, page 125.
M
Magnetic Shielding of Transformers at Audio Frequencies, W. G. Gustafson, page 416.
Maintenance, Telephone Circuit, New Transmission Measuring Systems for, F. H.
Best, page 1.
Meacham, L. A., The Bridge Stabilized Oscillator, page 574.
Mills, R. H., H. J. Fisher and M. L. Almquist, A New Single Channel Carrier Tele-
phone System, page 162.
Montgomery, H. C, An Optical Harmonic Analyzer, page 406.
Morgan, S. 0. and E. J. Murphy, The Dielectric Properties of Insulating Materials,
page 640.
Motion Picture Photography, High Speed, W. Herriott, page 393.
6
BELL SYSTEM TECHNICAL JOURNAL
Mumford, W. W., C. R. EnglundandA. B. Crawford, Ultra-Short-Wave Transmission
and Atmospheric Irregularities, page 489.
Murphy, E. J. and S. 0. Morgan, The Dielectric Properties of Insulating Materials,
page 640.
Murphy, 0. J. and A. C. Norwine, Characteristic Time Intervals in Telephonic
Conversation, page 281.
N
New Telephone Sets, Instruments for the, W. C. Jones, page 338.
New Telephone Sets, Transmission Features of the, A. H. Inglis, page 358.
Norwine, A. C, Devices for Controlling Amplitude Characteristics of Telephonic
Signals, page 539.
Norwine, A. C. and 0. J. Murphy, Characteristic Time Intervals in Telephonic
Conversation, page 281.
O
Oscillator, The Bridge Stabilized, L. A. Meacham, page 574.
Oscillators, Stabilized Feedback, G. H. Stevenson, page 458.
Rack, A. J., Effect of Space Charge and Transit Time on the Shot Noise in Diodes,
page 592.
Radio: Ultra-Short-Wave Transmission and Atmospheric Irregularities, C. R.
Englund, A. B. Crawford and W. W. Mumford, page 489.
Radioactivity — Artificial and Natural, Karl K. Darrow, page 292.
Research, Communication, Spectrochemical Analysis in, Beverly L. Clarke and A. E.
Ruehle, page 381.
Ruehle, A. E. and Beverly L. Clarke, Spectrochemical Analysis in Communication
Research, page 381.
S
Samuel, A. L., C. E. Fay and W. Shockley, On the Theory of Space Charge between
Parallel Plane Electrodes, page 49.
Schelkunoff, S. A., The Impedance Concept and its Application to Problems of
Reflection, Refraction, Shielding and Power Absorption, page 17.
Shockley, W., C. E. Fay and A. L. Samuel, On the Theory of Space Charge between
Parallel Plane Electrodes, page 49.
Short-Wave, Ultra-, Transmission and Atmospheric Irregularities, C. R. Englund,
A. B. Crawford and W. W. Mumford, page 489.
Shot Noise in Diodes, Effect of Space Charge and Transit Time on the, A. J. Rack,
page 592.
Space Charge between Parallel Plane Electrodes, On the Theory of, C. E. Fay, A. L.
Samuel and W. Shockley, page 49.
Space Charge and Transit Time on the Shot Noise in Diodes, Effect of, A. J. Rack,
page 592.
Spectrochemical Analysis in Communication Research, Beverly L. Clarke and A. E.
Ruehle, page 381.
Stevenson, G. H., Stabilized Feedback Oscillators, page 458.
Strieby, M. E., Coaxial Cable System for Television Transmission, page 438.
Telephonic Signals, Devices for Controlling Amplitude Characteristics of, A. C.
Norwine, page 539.
Teletypewriters Used in the Bell System, Fundamentals of, E. F. Watson, page 620.
Television Transmission, Coaxial Cable System for, M. E. Strieby, page 438.
Time Intervals in Telephonic Conversation, Characteristic, A. C. Norwine and 0. J.
Murphy, page 281.
Transformers at Audio Frequencies, Magnetic Shielding of, W. G. Gustafson, page 416.
Transmission Features of the New Telephone Sets, A. H. Inglis, page 358.
Transmission Line, The Exponential, Chas. R. Burrows, page 555.
7
BELL SYSTEM TECHNICAL JOURNAL
Transmission Measuring Systems for Telephone Circuit Maintenance, New, F. H.
Best, page 1.
Tucker, R. S., M. A. Weaver and P. S. Darnell, Crosstalk and Noise Features of
Cable Carrier Telephone System, page 137.
W
Watson, E. F., Fundamentals of Teletypewriters Used in the Bell System, page 620.
Weaver, M. A., R. S. Tucker and P. S. Darnell, Crosstalk and Noise Features of
Cable Carrier Telephone System, page 137.
Wire Spacing Problems, Telephone Line, Studies of, J. A. Carr and F. V. Haskell,
page 195.
Wright, S. B., Amplitude Range Control, page 520.
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