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THE BELL SYSTEM
TECHNICAL JOURNAL
A JOURNAL DEVOTED TO THE
SCIENTIFIC AND ENGINEERING
ASPECTS OF ELECTRICAL
COMMUNICATION
EDITORIAL BOARD
Banxroft Gherardi F. B. Jewett
H. P. Charlesworth W. H. Harrison E. H. Colpitts
L. F. Morehouse H. D. Arnold O. B. Blackwell
Philander Norton, Editor J. O. Perrine, Associate Editor
TABLE OF CONTENTS
AND
INDEX
VOLUME X
1931
AMERICAN TELEPHONE AND TELEGRAPH COMPANY
NEW YORK
£|riodi(»l
PRINTED IX U. S. A.
7ayi.-t7
J* /5 •32
THE BELL SYSTEM
TECHNICAL JOURNAL
VOLUME X, 1931
Table of Contents
January, 1931
The Detection of Two Modulated Waves which Differ SHghtly in
Carrier Frequency — Charles B. Aiken 1
A Magnetic Curve Tracer — F. E. Haivortli 20
A Multi-Channel Television Apparatus — Herbert E. Ives 33
Condenser and Carbon Microphones — Their Construction and Use
—W. C. Jones 46
Certain Factors Affecting the Gain of Directive Antennas —
G. C. Soiithworth 63
Absolute Calibration of Condenser Transmitters — L. J. S'wian ... 96
Rating the Transmission Performance of Telephone Circuits —
W. H. Martin 116
Paragutta, A New Insulating Material for Submarine Cables —
A. R. Kemp 132
April, 1931
Symposium on Coordination of Power and Telephone Plant
Introductory Remarks — R. F. Pack 155
I — Trends in Telephone and Power Practise as Affecting
Coordination — IV. H. Harrison and A. E. Silver ... 159
II — Status of Joint Development and Research on Noise Fre-
quency Induction — H. L. Wills and 0. B. Blackivell 184
III — Status of Joint Development and Research on Low-Fre-
quency Induction — R. N. Comvell and H. S. Warren 206
IV — Status of Cooperative Work on Joint Use of Poles —
/. C. Martin and H. L. Hither 231
Closing Remarks — B. Gherardi 241
Overseas Radio Extensions to Wire Telephone Networks —
Lloyd Espenschied and William Wilson 243
Some Optical Features in Two-Way Television — Herbert E. Ives 265
Bayes' Theorem : An Expository Presentation — Edimrd C. Molina 273
Extensions to the Theory and Design of Electric Wave-Filters —
Otto J. Zobel 284
DELL SYSTEM TECHNICAL JOURNAL
July, 1931
Some Physical Characteristics of Speech and Music —
Harvey Fletcher 349
The Statistical Energy-Frequency Spectrum of Random Disturb-
ances — John R. Carson 374
Bridge Methods for Locating Resistance Faults on Cable Wires —
T. C. Henneherger and P. G. Edwards 382
Mutual Impedance of Grounded Wires Lying on the Surface of the
Earth— i^onoW M. Foster 408
Transients in Grounded Wires Lying on the Earth's Surface —
John Riordan 420
Developments in the Manufacture of Lead-Covered Paper-In-
sulated Telephone Cable — John R. Shea 432
Effect of Ground Permeability on Ground Return Circuits —
W. Hozvard Wise 472
Negative Impedances and the Twin 21-Type Repeater — -
George Crisson 485
New Standard Specifications for \\'ood Poles — R. L. Jones 514
October, 1931
The Interconnection of Telephone Systems — Graded Alultiples —
R. I. Wilkinson 531
Moving Coil Telephone Receivers and Microphones —
E. C. Wente and A. L. Thiiras 565
Some Developments in Common Frequency Broadcasting —
G. D. Gillett S77
Application of Printing Telegraph to Long-Wave Radio Circuits —
A. Bailey and T. A. McCann 601
Audible Frequency Ranges of INTusic, Speech and Noise —
W. B. Snoiv 616
Contemporary Advances in Physics, XXII — Transmutation —
Karl K. Darroiv 628
Developments in Short-Wave Directive Antennas — E. Bruce 656
Index to Volume X
Aiken, Charles B., The Detection of Two Modulated Waves which Differ Slightly
in Carrier Frequency, page 1.
Antennas, Directive, Certain Factors Affecting the Gain of, G. C. Soiitlnvorth,
page 63.
Antennas, Short-Wave Directive, Developments in, E. Bnicc, page 656.
B
Bailey, A., and T. A. McCann, Application of Printing Telegraph to Long-Wave
Radio Circuits, page 601.
Bayes' Theorem : An Expository Presentation, Edzvard, C. Molina, page 273.
Blackwcll, 0. B. and H. L. Wills, Status of Joint Development and Research on
Noise Frequency Induction, page 184.
Broadcasting, Common Frequency, Some Developments in, G. D. Gillctt, page 577.
Bruce, E., Developments in Short-Wave Directive Antennas, page 656.
Cable Wires, Bridge Methods for Locating Resistance Faults on, T. C. Henne-
bergcr and P. G. Edwards, page 382.
Cables, Submarine — Paragutta, A New Insulating Material for, A. R. Kemp,
page 132.
Carbon, and Condenser, Microphones — Their Construction and Use, W. C. Jones,
page 46.
Carrier Frequency. The Detection of Two Modulated Waves which Differ Slightly
in, Charles B. Aiken, page 1.
Carson, John R., The Statistical Energy-Frequency Spectrum of Random Dis-
turbances, page 374.
Circuits, Telephone, Rating the Transmission Performance of, W. H. Martin,
page 116.
Condenser and Carbon Microphones — Their Construction and Use, W. C. Jones,
page 46.
Condenser Transmitters, Absolute Calibration of, L. J. Sivian, page 96.
Contemporary Advances in Physics, XXII — Transmutation, I'Carl K. Darrozv paee
628.
Conzvell, R. N. and H. S. Warren, Status of Joint Development and Research
on Low-Frequency Induction, page 206.
Coordination of Power and Telephone Plant, Symposium on, pages 155-241.
Coordination, Trends in Telephone and Power Practise as Affecting, W. H. Har-
rison and A. E. Silver, page 159.
Crisson, George, Negative Impedances and the Twin 21-Type Repeater, page 485.
Darrow, Karl K., Contemporary Advances in Physics, XXII — Transmutation page
628.
Detection of Two Modulated Waves which Differ Slightly in Carrier Frequency,
The, Charles B. Aiken, page 1.
BELL SYSTEM TECHNICAL JOURNAL
Developments in Common Frequency Broadcasting, Some, G. D. Gillctt, page 577.
Developments in Short-Wave Directive Antennas, E. Bnicc, page 656.
Directive Antennas, Certain Factors Affecting the Gain of, G. C. Soiitlnvorth,
page 63.
E
Edzi'ards, P. G. and T. C. Ilcnncbcrgcr, Bridge Methods for Locating Resistance
Faults on Cable Wires, page 382.
Espenschicd, Lloyd and IVilliam Wilson, Overseas Radio Extensions to Wire
Telephone Networks, page 243.
Faults, Resistance, on Cable Wires, Bridge Methods for Locating, T. C. Hcnne-
bcrgcr and P. G. Edzmrds, page 382.
Filters, Wave. Extensions to the Theory and Design of Electric, Otto J. Zobel
page 284.
Fletcher, Harvey, Some Physical Characteristics of Speech and Music, page 349.
Poster, Ronald M., Mutual Impedance of Grounded Wires Lying on the Surface
of the Earth, page 408.
Frequency, Carrier, The Detection of Two Modulated Waves which Differ Slightly
in, Charles B. Aiken, page \.
Frequency Broadcasting, Common, Some Developments in, G. D. Gillett, page 577.
Frequency Ranges of Music, Speech and Noise, Audible, W. B. Snoiv, page 616.
Frequency, Energy, Spectrum of Random Disturbances, The Statistical, John R.
Carson, page 374.
G
Gherardi. B., Closing Remarks (in the Symposium on Coordination of Power and
Telephone Plant), page 24L
Gillctt, G. D., Some Developments in Common Frequency Broadcasting, page 577.
H
Harrison, IV. H. and A. E. Sihcr, Trends in Telephone and Power Practise as
Affecting Coordination, page 159.
Hazvorth, P. E., A Magnetic Curve Tracer, page 20.
Hennchcrgcr and P. G. Edzvard.<;, Bridge Methods for Locating Resistance Faults
on Cable Wires, page 382.
Huber, H. L. and /. C. Martin, Status of Cooperative Work on Joint Use of Poles,
page 231.
Impedance of Grounded Wires Lying on the Surface of the Earth, Alutual, Ronald
M. Foster, page 408.
Impedances, Negative, and the Twin 21 -Type Repeater. George Crisson, page 485.
Induction, Low-Frequency. Status of Joint Development and Research on, R. N.
Conzvell and IF. S. Warren, page 206.
Induction, Noise Frequency, Status of Joint Development and Research on H L
Wills and O. B. Blaekzcell, page 184.
Interconnection of Telephone Systems, The— Graded Multiples, R. I. Wilkinson,
page 531.
Ives, Herbert E., A Multi-Channel Television Apparatus, page 33.
Ives, Herbert E., Some Optical Features in Two-Way Television, page 265.
BELL SYSTEM TECHNICAL JOURNAL
J
Jones, R. L., New Standard Specifications for Wood Poles, page 514.
Junes', W. C, Condenser and Carbon Microphones— Their Construction and Use,
page 46.
K
Kemp A R., Paragutta, A New Insulating Material for Submarine Cables, page
"132.
M
McCann, T. A. and A. Bailey. Application of Printing Telegraph to Long-Wave
Radio Circuits, page 601.
Magnetic Curve Tracer, A, F. E. Hazvorth, page 20.
Manufacture of Lead-Covered Paper-Insulated Telephone Cable. Developments in
the, John R. Shea, page 432.
Martin, J. C. and H. L. Hiiber, Status of Cooperative Work on Joint Use of Poles,
page 231.
Martin W H Rating the Transmission Performance of Telephone Circuits, page
116.
Microphones. Condenser and Carbon— Their Construction and Use, W. C. Jones,
page 46.
Microphones, Moving Coil Telephone Receivers and, E. C. JVente and A. L.
Thiiras, page 565.
Molina, Edivard C, Bayes' Theorem: An Expository Presentation, page 273.
Multiples, Graded— The Interconnection of Telephone Systems, R. I. Wilkinson,
page 531.
Music, Some Physical Characteristics of Speech and, Harvey Fletcher, page 349.
Music, Speech and Noise, Audible Frequency Ranges of, W. B. Snoiv, page 616.
N
Networks, Wire Telephone, Overseas Radio Extensions to, Lloyd Espenschied and
William Wilson, page 243.
Noise Frequency Induction. Status of Joint Development and Research on, H. L.
Wills and O. B. Blacku'ell, page 184.
Noise, Speech and Alusic, Audible Frequency Ranges of, W. B. Snozv. page 616.
O
Optical Features in Two-Way Television, Some, Herbert E. Ires, page 265.
P
Pack, R. F.. Introductory Remarks fin the Symposium on Coordination of Power
and Telephone Plant), page 155.
Paragutta, A New Insulating Material for Submarine Cables, A. R. Kemp, page
132.
Permeabilit}-, Ground, on Ground Return Circuits, Effect of, W. Hozcard IJ^ise,
page 472.
Physics. XXII, Contemporary Advances in — Transmutation, Karl K. Darrozv,
page 628.
Poles, Status of Cooperative Work on Joint Use of, /. C. Martin and H. L. Hiiher,
page 231.
Poles, Wood, New Standard Specifications for. R. L. Jones, page 514.
Power and Telephone Plant, Symposium on Coordination of. pages 155-241.
Printing Telegraph, Application of, to Long-Wave Radio Circuits, A. Bailey and
T. A. McCann, page 601.
7
BELL SYSTEM TECHNICAL JOURNAL
R
Radio Circuits, Long-Wave, Application of Printing Telegraph to, A. Bailey and
T. A. McCann, page 601.
Radio: The Detection of Two Modulated Waves which Differ Shghtly in Carrier
Frequency, Charles B. Aiken, page 1.
Radio : Developments in Short-Wave Directive Antennas, E. Bruce, page 656.
Radio: Some Developments in Common Frequency Broadcasting, G. D. Gillett,
page 577.
Radio Extensions to Wire Telephone Networks, Overseas, Lloyd Espenschied and
William Wilson, page 243.
Receivers and Microphones, Moving Coil Telephone, E. C. Wcnte and A. L.
Thiiras, page 565.
Repeater, Twin 21 -Type, Negative Impedances and the, George Crisson, page 485.
Riordan, John, Transients in Grounded Wires Lying on the Earth's Surface, page
420.
Shea, Jolin R., Developments in the Manufacture of Lead-Covered Paper-Insulated
Telephone Cable, page 432.
Silver, A. E. and W. H. Harrison, Trends in Telephone and Power Practise as
Afifecting Coordination, page 159.
Sivian, L. J., Absolute Calibration of Condenser Transmitters, page 96.
Snozv, W. B., Audible Frequency Ranges of Music, Speech and Noise, page 616.
Southworth, G. C, Certain Factors afifecting the Gain of Directive Antennas, page
63.
Specifications, New Standard, for Wood Poles, R. L. Jones, page 514.
Speech and Music, Some Physical Characteristics of, Harvey Fletcher, page 349.
Speech, Music and Noise, Audible Frequency Ranges of, W. B. Snozv, page 616.
Statistical Energy-Frequency Spectrum of Random Disturbances, John R. Carson,
page 374.
Submarine Cables — Paragutta, A New Insulating Material for, A. R. Kemp, page
132.
Telegraph, Printing, Application of to Long-Wave Radio Circuits, A. Bailey and
T. A. McCann, page 601.
Telephone Networks, Wire, Overseas Radio Extensions to, Lloyd Espenschied, and
William Wilson, page 243.
Television, Two-Way, Some Optical Features in, Herbert E. Ives, page 265.
Television Apparatus, A Multi-Channel, Herbert E. Ives, page 2)2.
Thuras, A. L. and E. C. Wente, Moving Coil Telephone Receivers and Micro-
phones, page 565.
Transients in Grounded Wires Lying on the Earth's Surface, Jolm Riordan, page
420.
Transmission Performance of Telephone Circuits, Rating the, W. H. Martin, page
116.
Transmitters, Condenser, Absolute Calibration of, L. J. Sivian, page 96.
W
Warren. H. S. and R. N. Conzvcll, Status of Joint Development and Research on
Low-Frequency Induction, page 206.
Wave-Filters, Electric, Extensions to the Theory and Design of, Otto J. Zobel,
page 284.
8
BELL SYSTEM TECHNICAL JOURNAL
Jl'cnte, E. C. and A. L. Thuras, Moving Coil Telephone Receivers and Micro-
phones, page 565.
]\'ilkinson, R. L, The Interconnection of Telcphdiie Sj'Stems — Graded Multiples,
page 531.
If ills, H. L. and O. B. Blackell, Status of Joint Development and Research on
Noise Frequency Induction, page 184.
ll'ilson, William and Lloyd Espcnschied, Overseas Radio Extensions to Wire Tele-
phone Networks, page 243.
Wire Telephone Networks, Overseas Radio Extensions to, Lloyd Espcnschied and
William Wilson, page 243.
]]'ise, W. Hozvard, Effect of Ground Permeabilitj- on Ground Return Circuits,
page 472.
Zobcl, Otto J., Extensions to the Theory and Design of Electric Wave-Filters,
page 284.
The Bell System Technical Journal
January, 1931
The Detection of Two Modulated Waves Which Differ
Slightly in Carrier Frequency *
By CHARLES B. AIKEN
The present paper coiUains an analysis ol the detection of two waves
modulated with the same, or with different, audio frequencies and differing
in carrier frequency by several cycles or more. Both parabolic and straight
line detectors are treated and there are derived the expressions for all of the
important audio frequencies present in the output of these detectors when
such waves are impressed. There are discussed the types of interference
which result when one station is considerably weaker than the other and
simple attenuation formulae are employed in estimating the character and
extent of the interference areas around the two transmitters. Beyond the
use of such formulae no attention is given to phenomena which may occur in
the space medium such as fading, diurnal variations in field intensity, etc.
WHENEVER one of two stations operating on the same wave-
length assignment wanders from its proper frequency, waves
are Hkely to be received which differ in carrier frequency by several
cycles or more. Ihider such conditions the two signals may be thought
of as made up of entirely distinct frequencies and phase relations
lietween analogous components of the two waves need not be con-
sidered. In the important case in which the carriers are of identical
frequency this is no longer true and phase and its dependence on
position and transmission phenomena must be taken into account.
This case will be reserved for future study, the present work being
limited to a consideration of the phenomena connected with the
detection of distinct frequencies.
The most important undesired frequency which is present in
the output of the detector is the beat note between the two carriers.
It is sometimes carelessly assumed that if the frequency of this beat
note is reduced below the audible range the only remaining interference
will be due to the speech from the undesired station. Such is not the
case and it will be shown later on that when the beat frequency is
reduced below the audible range, but not to zero, there remains a group
of spurious frequencies which will introduce an interfering background.
When the undesired carrier is of relatively small intensity this back-
ground is a great deal stronger than the interfering speech. It is
therefore desirable to obtain quantitative data on the interfering spec-
* Proc, I. R. E., Jan., 1931.
1
2 BELL SYSTEM TECHNICAL JOURNAL
trum which occurs in the receiver output, in terms of the intensities
and degrees of modulation of the input signals.
It is to be expected that the results obtained will depend, to some
extent at least, on the type of detector which is used. The square law^
characteristic is a fair approximation to that of any detecting device
which is worked over only a small range and hence an analysis of this
characteristic may be expected to serve as an excellent guide to general
detector performance. When large signals are impressed on the
detector the functioning of the device may approximate more closely
to that of the ideal straight line detector. It has been felt that a study
of these two types would furnish data from which the performance of
any intermediate type of detector could be inferred without great error.
As the problem of the square law detector is very nmch the simpler it
will be considered first.
Mathematical Analysis
There will be assumed two broadcasting stations transmitting on
frequencies which differ by a relatively small amount, the beat fre-
quency being restricted to the audible range or less. Each of the
carriers will be assumed to be modulated by a single audio frequency,
the modulating frequencies at the two stations being, in general,
different. The total signal impressed on the receiving detector will
then be of the form
V = £(1 + -1/ cos pt) cos (joit + c{\ -\- m cos qt) cos oi-J, (1)
in which
V is the total alternating voltage impressed on the detector.
E is the amplitude of the desired carrier.
e is the amplitude of the undesired carrier.
M is the degree of modulation of the desired signal.
m is the degree of modulation of the undesired signal.
a;i/27r is the frequency of the desired carrier.
co2/27r is the frequency of the undesired carrier.
pjlir is the frequency of the desired modulation.
5/2 TT is the frequency of the undesired modulation.
Square Law Detector
We shall first suppose this signal to be impressed on a detector which
will be assumed to have a characteristic in the neighborhood of the
operating point, of the form
i = Ao + A,v+A,v\ (2)
DETECTION OF TWO MODULATED WAVES
An expression of this type will accurately represent a small portion of
any continuous characteristic. The present analysis requires that the
impressed e.m.f. shall be of small amplitude in order that the limits of
the portion of the characteristic thus represented may not be exceeded.
This restriction is necessary in treating square law detectors.
The audio frequency output of the detector will be due entirely to
the second order term in (2). Hence it will be sufificient, for our
purposes, to square the expression for v. We are interested primarily
in the ratios of the amplitudes of the various undesired audio fre-
quencies produced to the amplitude of the desired signal of frequency
Pi lit. Such a ratio will be designated as a relative amplitude. Neg-
lecting circuit constants, etc., which will apply equally in all the
expressions for the various frequencies, the amplitude of the desired
component of the audio frequency output is readily shown to be E-M.
The expression for v- is reduced to first power sinusoids and the ampli-
tude of each frequency converted to a relative amplitude by dividing
by E~M. The case in hand >ields twelve undesired audio frequencies,
the relative amplitudes of which are listed in table I. Before com-
menting on these results we shall consider the straight line detector.
TABLE I
Angular
Velocity
Ratio to
Angular
Velocity
Ratio to
Em
IP
M
i
p ± U
e
2E
e-in
EHI
q zizU
em
2
2EM
2q
â– iET-M
e
p ±g. ± u
em
51
u
EM
in which 11 = 0:1 — coo.
The Straight Line Detector
In making analyses of rectification by a straight line detector it is
customary to reduce the sum of the various impressed radio frequencies
to a single radio frequency, the amplitude and phase angle of which are
slow functions of time. The most common example of this type of
treatment is a combination of the carrier and two side bands of single
frequency modulation into the familiar expression for a modulated
4 BELL SYSTEM TECHNICAL JOURNAL
wave in which tlie aniphLude of the radio frequency is an audio
frequency function. In this case the radio frequency phase angle is
constant. In the case of a single frequency modulation with one side-
band eliminated there are impressed on the detector input only two
frequencies. These may be combined in a well known manner. ■•
Thus, if the impressed voltages are of the form acosx and bcosy, then
the amplitude is given by
^la- + b'^ + lab cos (.r - y). (3)
The expression for the phase angle will not be given here as it can be
shown that if a and b are unequal and the difference between the
frequencies xllir and yjlir is small compared with either frequency,
then the variation of the phase angle with time may be neglected in
computing the audio frequency components. In the present case we
have two radio frequency waves the amplitudes of which are not
constants but are slow functions of time and these may be substituted
for a and b in (3). Thus the effective amplitude of the total input
signal may be taken to be
5 - V^- + B' -\- 2AB cos Id, (4)
in which
A = E{\ + AI cos pt),
B = e{l + m cos qt),
and
U — 0)1 — C02.
The problem then resolves itself into an analysis of the detection, by a
straight line detector, of a single radio frequency component. The
results of such an analysis are well known and it can be readily shown
that the audio frequency output may be obtained, except for a factor
of proportionality, by resolving the amplitude into its audio frequency
components. In the present case the amplitude to be resolved is
given by (4) which may be written
S = V(^ + B)^ - 2AB{\ - cos ut).
The interfering signal B will be taken to be always less than the de-
sired signal A, and hence A- -\- B- > 2AB, from which it follows that
{A + By > 2AB{1 — cos «/.) Hence the radical may be expanded
by the binominal theorem, giving
.45(1 - cos ut)
S = A -\- B -
A + B
A'^B~[\ - cos Kty- A'B\\ - cos ut)
2 (.4 + By 2iA + By
^ \'ide: Lord Rayleigh, "Theory of Sound," i)age 23, sec. ed.
C^)
DETECTION OF TWO MODULATED ir.H7-:.S 5
It is to be observed thai eacli of I lie lernis of this series, except the
first, contains time in the denominator and hence further expansions
are necessary. The denominators of the various terms can be ex-
panded by the binominal theorem in such a way as to put all the
expressions containing time in the numerators, the expansions being in
powers of
{ME cos pt + »J& cos qt)l{E + e).
By the proper trigonometric transformations it is possible to reduce the
final expression for S to frequencies in p, q, u and the sums and differ-
ences of the various multiples of these quantities. An additional
discussion of this analysis is given in an appendix. In order that the
various series involved may converge with a manageable degree of
rapidity it is necessary to limit the relative amplitudes of the interfering
carriers and the degrees of modulation as well. Consecjuently the
solutions are restricted to intensities of the interfering carrier of 0.1, or
less, of the desired carrier and to degrees of modulation of either signal
ranging from 0.1 to .5. These limits are suitable also because we are
interested chiefly in interference by a relatively weak signal, the inter-
ference caused by a signal, the carrier amplitude of which is greater
than 0.1 of that of the desired carrier amplitude being near the tolerable
limit in the majority of cases. The upper value for the modulation of
0.5 is approximately equal to the average degree of modulation of a
station employing as deep modulation as is practical, only the peaks
running up to nearly unity. The value of 0.1 for the lower limit is of
course transgressed by soft passages in speech or music. However, the
range here specified is sufficiently large to give an excellent idea of
what may be expected from various degrees of modulation of desired
and interfering signals and the results of more extreme cases may be
inferred from the data here developed. Under these limits it is found
that the only audio frequencies of any importance which appear in
the output are:
5 = ( ME - eg ia,M - ai + a,^\ + '"''"^.^^' ) cos pt
, / / ciiMm meg\ .^e~fboni\
+ ( me - egi aom ^ ) jj^ — ) cos qi
, / / auM m-eg\ h^e^g^ , i ox \ ,
+ [eg (go ^ Y^ ] + -j^ (2 + nr) j cos ut
7^ cos 2ui
BELL SYSTEM TECHNICAL JOURNAL
I / (inH! (uMm me^\ io^V^ \ / , ^, /-x
In which
flo = 1 + -^ + — g^ . ai = Mg + -^ + — g^
flo = ^ + ^, 6o = 1 + 3.1/2g2^ ^ (7a)
Comparison Between Detectors
It is now possible to make a comparison between the performance
of the straight Hne and the square law detectors. In Figs. 1 to 4 are
shown the relative amplitudes of the interfering frequencies in the two
cases for various degrees of modulation. The data for the square law
case are indicated by dashed lines and for the straight line case by
solid lines, and where the two coincide this is noted on the figures. It
is to be noted that the expression for the amplitude of the desired
frequency P/2t is a complicated function. However, computation
shows that over the range in which we are interested, the value of this
expression does not differ from ME by more than 1 per cent and,
therefore, this value has been assumed in computing the relative
amplitudes of the other frequencies.
Probably the most striking feature to be noted in comparing the two
cases is the similarity of the results. This is particularly evidenced by
the carrier beat note of frequency u/2t the amplitude of which differs
in the two cases by an inappreciable amount. The spurious fre-
quencies (q ± n)/2Tr also are practically identical for both detectors.
There are, howe\er, several important differences as follows:
The group of spurious frequencies of angular velocity p ± q ± u,
which is of appreciable importance in the square law case, is entirely
absent from the range of magnitude considered when a straight line
detector is employed. The frequencies {p ± u)l2Tr are greater in the
square law case over the range which we have considered, but the
curve which represents them has a smaller slope than in the straight
line case and for larger values of the interfering signal the intensities of
these frequencies would be relatively less with the square law detector.
The intensity of the undesired speech q is definitely less in the straight
line case than in the square law case but the slope of the q curves is
DETECTION OF TWO MODULATED WAVES 7
about the same for both except for M = m = 0.5. It is of interest to
observe that the interfering speech received on the straight line detector
is verv much less in intensity than would be the case if the strong
desired signal were absent, and that the variation of the amplitude of
this frequency with intensity of the undesired carrier is greater when
the desired frequency is present. We have here an analytical descrip-
tion of the familiar masking effect which occurs when a strong unmodu-
lated carrier is received simultaneously with a weak modulated signal.
For example, when e/E = 0.1 it can be seen from Fig. 1 that the
relative amplitude of the component of frequency g/lTr is 0.0063 for
the case of the straight line detector. If this component were un-
affected by the presence of the strong signal it would have an amplitude
proportional to em and a relative amplitude of em/EM which for the
values here considered is 0.1. Hence the "masking" effect is here
responsible for a reduction of 24 db.
Lastly, it may be mentioned that there are in the case of the straight
line detector certain frequencies of small amplitude which are entirely
absent from the square law case. However, no frequency is shown
the relative amplitude of which is less than 0.01 for all four pairs of
values of M and m, as such frequencies are unimportant. An exception
is made with regard to ^ ± u. This is always less than 0.01 over the
range considered but is included for the sake of comparison with the
square law results.
Further Coxsideratiox of Detector Output
The second harmonic of the desired signal is of importance only in
the square law case. It is of the nature of a distortion which is inde-
pendent of the interference and may be omitted from the consideration
of the undesired audio frequencies which are a result of the interference.
From Figs. 1 to 4 it is evident that the most important interfering
frequencies are those of angular velocity, u, q ±n, p ± u and p ± q_
± 11, the last being of importance only in the case of the square law
detector. It is with these frequencies, together with that of the
interfering speech qjlir, that we shall be chiefly concerned.
When the relative magnitudes of the interfering frequencies, which
are tabulated on page 3, are multiplied by E'^M, the resulting quantities
are proportional to the absolute magnitudes of these frequencies. It
is to be noted that the frequencies of greatest interest have absolute
magnitudes which are linear functions of J/ or m except {p ±q ± u)l2-ir
which is proportional to niM, and 21 /It which is independent of both M
and 7)1 and will, therefore, be unaffected by the type of modulation
employed at either station. In case there are several frequencies
8 BELL SYSTEM TECHNICAL JOURNAL
present in the modulation of each station the radio frequency waves
will be of the form E{\ + J/j cos pit + -1/2 cos p2t -{- • • • ) cos wi/ and
e(l + 7ni cos qit + m^ cos qit + • • •) cos uoi- For every frequency of
the former case which contained .1/ as a factor of its amplitude we shall
now have several frequencies respectively proportional to Mi, Mo etc.
while an analogous new group will correspond to the former frequencies
.01
.001
.0001
—
1 J
\ /
-o>^ -
DESIRED SIGNAL^ E +MCOSPT) COS w, T
UNDESIRED SlGNAL= e (l +mCOSQT) COSu^sT
/\
/
/
DATA FOR SQUARE LAW DETECTOR
— _
/
^
/
M-.i m=.l
~<^jf
/
f
^
-^y
/
r?>
^
.,/
/
M
— -
2P_
--
-
--
â– /â–
-
—
—
—
Z^
—
/
/
/
^ t
/
t J
^
Oif
Ltr
/
'iV
/
/
/
f' J
/
.0/
'X
/
^\f/l
/
^
>>
f
f
/
h
/
/
/
1 ^^
/
/
/
4
r
}
.001
.01
T=RATIO OF CARRIER AMPLITUDES
Fig. 1 — Relative am])litiides of undesired frequencies as a function of the ratio of
the amplitudes of the desired and the interfering carriers. .Modulation of both
stations small and equal.
containing m. Hence we shall have two frequency spectra derived
from the desired speech spectrum containing the ^'s, but one of the
spectra will be shifted upward in frequency by an amount ujlir and
the other downward by the same amount. Two additional spectra
will be derixed in a similar manner from the undesired speech spectrum
containing the (/'s. The frequencies of the type {p ± q ± u)'lTr\\\\\he.
DETECTION OF TWO MODULATED WAVES 9
numerous as there will be a product of the .l/'s with each of the ms.
However, these are of even moderate importance only when the
modulations of both stations are high, and a square law detector is
emplo\'ed at the receiver.
Hence we may picture the interference as made up chiefly of dis-
placed frequency spectra of the type mentioned above, of a carrier
.001
-=i+z^
M-.i m=.
s
^Jf-
7 --,
z y
/
4
/
/ /
/ 7^
>
1 /
/
/
\y
f—7^
-7^
^-
.' 1
/
J
cjy^
/ /
/
/ /
â– ' /
''/
-
2P
/â–
. .._
-/—
-7'-ki
~y /
/
/
/
/
/
■■—
/
<
If/
^'--^^
b
^
y
1
I'i'-i
z'
^r^^^
/
I-,
ti W-
/
:â– ^
/
/
4
7 ^
-f- — 1-1--
/ .
/ W
4
V
/
Z
/
/
•
A
/
/
/,
7.
5^^
/
'
/
'/
/
> - -
/
/
/
f f
J^_J
.0001
.001 -01
/erratic of carrier amplitudes
Fig. 2 — Relative amplitudes of undesired frequencies as a function of the ratio
of the amplitudes of the desired and the interfering carriers. Modulation of desired
station small and of interfering station large.
beat and of the interfering speech, which is weak but important because
of its intelligibility. The results in the case of a straight line detector
would not be very greatly different. The frequencies of the type
(p db 2 zt u)lliv would be negligible, the two spectra derived from
p ± u would be much less important and certain new, but rather small
cross product frequencies would appear.
10
BELL SYSTEM TECHNICAL JOURNAL
In estimating the interference the carrier beat can be considered by
itself and from the data at hand there can be derived the areas around
each of two stations having approximately the same carrier frequency,
inside of which the amplitude of the beat note will be down a given
number of db from that of the desired speech. The same is true of the
interfering speech when it is different from the desired speech. The
1.0
.0011
.0001
1
—
1
M=.5 nn=.i
— -
2£
-
.._
- —
-
—
A-
/
/
y
/
/
/
7
/
f
/
•
/
<
/
/
/
/
>
â– ^z
/
- A
«]
y
A'' -
>o/
/
-Q^ -
^^
/
-X -.
o9^
/
Z .?
/^
/
/
1
/
/
/
/''/
/
/
/
/
z
/
/
/
/
A
I
w
%-
.001 .01
RATIO OF CARRIER AMPLITUDES
Fig. 3 — -Relative amplitudes of undesired frequencies as a function of the ratio
of the amplitudes of the desired and the interfering carriers. Modulation of desired
station large and of interfering station small.
frequencies {p i zO'^Tr, (g ± iC)11-k. (p±qdz u)'2ir, etc., will coml>ine
to form a disturbing background which we shall designate as "displaced
side band interference." This may be taken to include all of the
interfering frequencies except those of the undesired speech and its
entirely unimportant harmonics. (The frequency 2p'2ir is not here
classed as an interfering frequenc}'.)
DETECTION OF TWO MODULATED WAVES
11
From Figs. 1 and 4 it is to be noted that when m = M the frequencies
{q rb u)l2-K are the largest components of the displaced side band
interference if a straight line detector is used and have the same
amplitude as the {p ± «)/27r components if a square law detector is
used. When m > M the q ± u group is much more important than
the p dz u group as is evident from Fig. 2. When M > m the 5 rh w
10
.001
N
A'.b m=.
•s
-A
-
--
2P
—
—
—
— f
â– ^^^^
"
/
7^
/
/\ 1 \^\
V
^
/
/
/
f
7
A
-7' 7^
V
^
y
v^
r-i-
,/
/
/
2 _,L_^
/
/
^<'
/
L-t-
/
0^
/
^-^
/
/
#f[
4
y
-f-fAr4--\-
/
r
/
/
/
f
1
/ y of
.0001
.001 .01
/^= RATIO OF CARRIER AMPLITUDES
Fig. 4 — Relative amplitudes of undesired frequencies as a function of the ratio of
the amplitudes of the desired and the interfering carriers. Modulation of both
stations large and equal.
group is less important but this case is of no great interest for if the
stations are transmitting identical programs, with similar degrees of
modulation, it cannot occur and if the programs are different then the
interference is determined primarily by what happens when m > M.
Consequently we may consider that the q ± u group constitutes the
most important part of the displaced side band interference except
12 BRI.L SYSTEM TF.CIINICAL JOURXAL
when a scjuare law detector is used and the })rograms are identical. In
such a case we shall assume tiiat both stations employ the same degree
of modulation and that therefore the q ± Ji and p ± 7i groups are of
the same im[)ortance.
Interference Areas of Stations
W'e have distinguished between three types of interference, namely,
carrier beat, unwanted speech and displaced side band. We shall now
compute, for several values of attenuation, percentage modulation
etc., the areas around a transmitting station inside of which each of
these types of interference, due to a second station, will have a relative
importance which is not greater than a certain specified amount.
In estimating these areas we must deal with two possible cases which
may arise in practice: (1) The two stations transmit different programs.
(2) The programs are the same. The carriers are assumed to differ in
frequency in both cases.
Case 1
The importance of the various types of interference which are
present, will be determined by their ratios to the intensity of the
desired speech. In the present case in which the two stations transmit
different programs, the amount of interference which may be tolerable
will be determined by what occurs when the modulation of the desired
station is low, while that of the interfering station is high. Hence, in
studying this case we shall make use of Fig. 2, which gives data com-
puted on the basis of a modulation of 0.1 for the desired station and
0.5 for the interfering station.
Taking up first the consideration of the carrier beat note, we shall
determine the cur\e along which the intensity of the beat is down a
given numl)er of db from the desired speech. The position of this
curve will depend on the degree of modulation of the desired signal,
since the lower the modulation the more noticeable \\\\\ be a beat note
of a given intensity. When we have specified the db difference which
must exist between these two components of the receiver output the
carrier ratio can be picked off' from the n line of Fig. 2.
In order to determine the curve along which this carrier ratio exists
we shall proceed as follows:
The desired station will be considered to be at the origin of a system
of rectangular coordinates and the undesired station will be at the
point {D, O). We shall assume that the powers of the desired and
undesired stations are Pi and P-i, respectively, and that their distances
from a point in the coordinate plane are di and J2; then if we denote the
DETECTION OF TWO MODULATED WAVES 13
ratio of the carriers by K = e/E the equation of the curve along which
the value of K is constant is given by:
^ ,-, = f:%-..,. (8)
This equation is based upon a convenient form of the Austin-Cohen ^
formula for the intensity of the field radiated from a radio transmitter.
This formula is:
in which X is the wave-length in meters, d is the distance from the
transmitter in miles and a is an attenuation constant which may range
from zero up to 0.01 or even more. In writing down equation (8) we
have used the abbreviation:
101. Sa^i ,.^x
From (8) there have been computed curves for the case in which
Pi = P2 and for various values of K and a. X has been taken as 300
meters and D, the distance between the stations, as 1,000 miles.
In Fig. 5 are shown several curves for a = 0.001. For small values
of K, the curves are practically circular and are of small area. As K
increases, the curves become oval shaped and it can be readily shown
that for values of K greater than a certain critical amount, the curves
will not close but will be of a shape which is roughly hyperbolic.
In Fig. 6 are shown curves corresponding to a value for a of 0.002.
It is to be noted that an increase in a enormously increases the area
inside of which the ratio of the carriers is less than a certain value.
The effect of a will of course be dependent upon the magnitude of the
distance between the stations and will be more pronounced the larger
this distance. For the present case in which D = 1,000 miles, there
is not much point in considering values of a larger than 0.002, since the
attenuation would be so great as to make the effect of one station on the
service area of the other of very little consequence.
If we specify that the carrier beat must be at least 40 db down from
the speech output due to a 10 per cent modulated signal, then curve 1 of
Figs. 5 and 6 will represent the areas inside of which this requirement
will be met, while if we call for an interval of 20 db between these two
components, curve 5 of Figs. 5 and 6 will represent the areas in which the
condition is satisfied. It is evident that if a rigid restriction is placed
on the permissible beat note interference which may be allowed, and if
the attenuation is of a small value then the area in which the beat
- L. VV. Austin, Proc, I. R. E., Vol. 14, p. 377.
14 BELL SYSTEM TECHNICAL JOURXAL
nole ma>' be neglected is extremely small. On the other hand this
area increases very rapidly as the attenuation increases.
We may use the same sets of curves in considering the displaced
side band interference. From Fig. 2 it is evident that by far the most
important components of this interference are those represented by the
{q ± ii) group. In order to estimate this interference we must follow
some rule for coml)ining the q -\- n component with the q — u com-
ponent. In order to do this in a strictly correct manner we should
have to take into account the frequencies and sensation levels of the
components. However, it has been shown ' that over a considerable
portion of the audio frequency range, and for sensation levels of
approximateh' the magnitude in which we are interested, the inter-
fering effect of these frequencies may be taken to be approximately
equal to that due to a single frequency of twice the amplitude of
either component. We shall therefore take our data from the dash-dot
curve of Fig. 2. h'rom this curve it appears that if the displaced side
band interference is to be 40 db down from the desired speech, we must
have a carrier ratio of 0.002, while if it is to be 20 db down from the
desired speech the corresponding carrier ratio is 0.02. The curves
corresponding to these values are shown by 2 and 6, respectively, on
Figs. 5 and 6.
From this it appears that the area in which the side band noise is not
objectionable may be a great deal larger than that in which the carrier
beat is of a tolerable intensity. If the frequency of the carrier beat is
reduced below the useful audible range then the former area may be
considered to be entirely free from interference of any kind. Conse-
quently, it is highly desirable to limit the maximum possible differences
in the carrier frequencies to a value which is definitely below the audio
frequency pass band of commercial radio receivers and loud speakers.
Turning now to the undesired speech, we note that it is of very little
importance compared with the displaced side band interference. Thus,
if this speech is to be 40 db down from the desired speech, the value of
the carrier ratio is 0.044 for the case of a square law detector, while for a
difference in level of 20 db, the carrier ratio is 0.14. A curve for the
case of a 40 db difference is indicated by 7 of F"ig. 5.
The comparison between curves 7 and 6 emphasizes the fact that we
may have considerable areas of intolerable displaced side band inter-
ference in which the intelligible speech from the undesired station is
not noticeable. Of course, this interference is often classed as distorted
speech but the distinction is convenient in the present discussion.
^ J. C. Steinberg, "The Relation Between the Loudness of a Sound and its Physical
Stimulus," Phys. Rev., Sec. Ser., Vol. 26, pp. 507-523.
DETECTION OF TWO MODULATED WAVES
Case 2
15
In this case the programs are identical and consequently the speech
from the two stations will undergo simultaneous fluctuations of
intensity. We shall here assume that the two stations have the same
degree of modulation at any instant. We may then take our data
from the curves for which M = m. However, this does not apply to
the carrier beat note, since its intensity is independent of the degree of
I
1
s
'
[
/
\,
1
/
\
\
/
K=.C
)06
N
/
N
\
\
/
fi
J04^
\
/
/
\
\
. DESIRED^
\STATION
,
A-i
J02\
'k=.OOI y
/
j/-
\ 5\4
aUft,
>w
:/
/
V
^
-^
\
\
h^
^
— ^
y.
/
4
/
/
y
y
\
— ^
\
\
.^7
.^-â– 02^^
. — â–
/
y
^
^
^
â– ^
^^^
-^
K-.l
^
^^
K«.2
^ .
r^
1
1
1000 1
i/1ILES
a(. =.001
UNDESIRED
STATION
Fig. 5 — Curves along which the ratio of the carrier aniplitudes received from two
stations has a constant value K, as indicated. Attenuation small.
modulation of either station and its interfering effect will be determined
by conditions which exist when the desired station has a low degree of
modulation. Hence the discussion of this component of the inter-
ference will be exactly the same as in the preceding case.
Referring to Figs. 1 and 4, it is evident that by far the greatest
portion of the displaced side band interference is due to the g ± w
components, in the case of the straight line detector, and the q ± u
and p ±^i components in the case of the square law detector. The
16
BELL SYSTEM TECHNICAL JOURNAL
identity of the curves for these components in the two figures shows
that the degree of modulation has practically no effect on the relative
importance of the interference which occurs when the same programs
are transmitted.
If we again assume that the total interference may be represented by
a fictitious component of twice the amplitude of the q -\- u component,
\
\
/
/
\
\
/
/
\,
\
\
/
/
/
A
\.
\
\
V
1
V
/
/
>.
\
\
v\
DESIRED
STATION
/
/
/
^
\
s\
^
/
y
^
^
X
.0
_ K».
— K =
5S^
2CA.-
^
^
^
^
'^
7
01
02
044 -
'^
^
^
^
â–
,
K = .l
. 9
%z
1
1000
»1ILES
c< =.002
UNDESIRED
STATION
Fig. 6 — Relative amplitudes of undesired frequencies as a function of the ratio of
the amplitudes of the desired and the interfering carriers. Attenuation constant
a twice that of Fig. 5.
we may take our data from the dash-dot line of Fig. 4. This should
represent the case fairly well for the straight line detector but when a
square law detector is used, greater interference should result due to the
importance of the p ± u terms. However, we shall consider only the
q zL u group and the phenomena associated with the square law case
may be readily inferred. In order that the displaced side band
interference may be 40 db down from the desired speech the carrier
ratio must have a value of 0.01, while if it is to be 20 db down, this
value must be 0.1. The first value corresponds to curves 5 of Figs.
DETECTION OF TWO MODULATED WAVES 17
5 and 6, while the second value corresponds to curves 8. We observe
that there is a tremendous difference between the areas which may be
considered to be free from displaced side band interference and those
which will be free from carrier l)eat interference, in case the beat
frequency is allowed to wander into the audible range. The comparison
between the two areas is given by curves 1 and 5 for the 40 db interval
and by curves 5 and 8 for the 20 db interval.
The speech from the interfering station will now be the same as the
desired speech and can have effect only in so far as it adds to or subtracts
from the desired speech. It will be noted from F'igs. 1 and 5 that for
carrier ratios of less than 0.1 this component is always down more than
40 db and may be safely neglected.
The foregoing discussion serves to illustrate the types of interference
which may be expected when two stations are operated on approxi-
mately the same frequency. The data discussed have involved low
values of attenuation. This is of particular interest when the distance
between stations is large since with high values of attenuation either
station will have very little effect on the service area of the other. Of
course at night time we may have signal strengths which will be of the
order of magnitude of that given by the simple inverse distance kiw
invoU'ing zero attenuation. This possibility probably presents a
serious limitation on night time common frequency broadcasting but
should be of little consequence during the daylight hours. Conditions
will be somewhat different for stations that are placed nearer together
and specific results can be readily computed for any given spacing.
The equations which have been discussed can be applied to any such
case and the areas corresponding to those in Figs. 5 and 6 determined.
One point which is emphasized by the results which have been
obtained is, that with a carrier frequency difference of several cycles
satisfactory reception cannot be expected in the regions which lie
midway between two transmitters. The field strength of one station
must be at all times predominately higher than that of the other and
consequently the use of pseudocommon frequency broadcasting should
be restricted to stations of wide geographic separation. It should then
be possible to furnish high grade service to relatively small densely
populated areas in the immediate vicinity of either transmitter,
reception at a considerable distance from both stations being ad-
mittedly unsatisfactory. However, if the carriers are strictly isoch-
ronous much larger service areas should be feasible.
IS BRLL SYSTEM TECHXICAL JOURNAL
AlM'KNDIX
Equation (5) is
I II
.IB(\ - cos ul)
S = A -\- B
A -^ B
III IV
y4 2^2(1 _ cos utY ylVi'(l - cos ut)
2{A -{- By 2iA -\- By
To expand tliese terms we write
1 1
{A -{-By (E -\- e -\- ME COS pt-\- me COS qty
1 / 11 {ME cos pt + me cos qt)
C^)
{E + ey \ E^e
}i(ii-\-\){ME cos pt-{-})ie co^qt)^
, , ,, v(n^\)(tJ + 2)---(}i-\-r-\)( ME cos pt-\-nie cos qt)'\ .. .
It is evident there are present in S an infinite number of frequencies
and it is necessary to select those which are of appreciable magnitude
relative to that of the desired frequency of amplitude £.1/. Fortu-
nately these are not very numerous.
In deciding whether or not a given term should be retained there
are two points to be considered: (1) whether all the terms of a given
frequency total to a value sufficiently large to call for the presence of
this term in the final result; (2) what per cent accuracy should be
required in the frequencies which are retained. Thus if it is desired to
retain all frequencies the relative amplitude of which is greater than
0.01 we cannot arbitrarily retain all individual terms which make a
contribution of 0.01 or greater and neglect those of relative importance
of less than 0.01. Thus if a term of a given frequency has a relative
amplitude of 0.01 and another term of the same frequency a relative
amplitude of 0.009 the second term should be retained. Otherwise we
should have a large percentage error in the value of the amplitude of
this frequency. On the other hand it is not desirable to maintain
the same degree of accuracy for the case of retained frequencies of
slight relative importance as for those of large importance. As a
compromise all individual terms have been retained which, after
division by EM, are of a magnitude greater than 0.005 for any values
of M , m and e'lE which are here dealt with. An exception is made in
DKTKCTIOX OF TWO MOIH'LATEP WAVES 19
the case of a term in cos pi derived from term III of (5). This term is
sHghtly larger than the above Hmit when M = 0.5 and e/E = 0.1 but
as it decreases rapidK with a decrease in e E it has been omitted for the
sake of simpHcity.
Having chosen this Hmit of 0.005 for the relative magnitude of
individual terms it can be shown to be permissible to neglect term IV
and all subsequent terms of (5). Furthermore, only a few of the large
number of terms yielded by III need be retained.
After applying these rules there appear several frequencies that are
never as large as 0.01 in relative magnitude and these ha\e been
omitted from consideration. As has been stated in the body of the
paper, an exception is made in the case of the frequencies {p±q±ii)/2ir.
If a given frequency exceeds 0.01 for any one of the four pairs of
values of .1/ and /;/, it has been shown on* the figures for all of the
pairs.
After the formula (5a) has been applied to 6' and the expressions for
A and B inserted there remains the necessity of reducing products and
powers of various sinusoidal terms to sums of simple first order
sinusoids. This is a tedious procedure but is a matter of simple
trigonometry and will not be set forth in detail.
From (5a) it can be seen that if M or m is near unity the series will
converge very slowh. Furthermore, since to obtain relative magni-
tudes we divide by M, it is impossible to obtain satisfactory convergence
due to small values of M in the denominator. Hence it is necessary to
limit M and m to 0.5 or less and in addition .1/ must be no smaller than
0.1. It would be permissible to allow m to become less than 0.1 but as
little would be gained by this ;;/ has been restricted to the same range
as AL
A Magnetic Curve Tracer
By F. E. HAWORTH
An apparatus for pliotograpliicall>' recording hysteresis loops and initial
magnetization curves is described. It employs a rotating drum and a tUix-
nieter, the restoring torque of the latter being completely counter-balanced
by a photoelectric cell arrangement. With this apparatus curves may be
taken so slowly that edfly currents are negligible. The accuracy of the
instrument is intrinsically as great as that of a ballistic galvanometer. An
anaKsis of sources of error is included.
FOR accurate determinations of hysteresis loops and initial magneti-
zation curves of magnetic specimens, a laborious routine involving
the use of a ballistic galvanometer is usually necessary. This article
describes an apparatus by means of which these curves may be obtained
photographically with quantitative accuracy. Attempts to devise
such a scheme have previously been made. Ewing ' describes one
which was used with short, thick specimens in a magnetic yolk.
Fleming^ invented a device, the Campograph, which made use of a
magnetometer and had the advantage of making possible the use of
long, thin, specimens, thus reducing eddy current and demagnetization
effects. J. B. Johnson ^ describes the most recently published design,
embodying a vacuum tube amplifier and a Braun tube oscillograph.
This hysteresigraph is used with frequencies of the order of five cycles
per second, or higher, and consequently introduces an eddy current
loss, a disadvantage in a great many measurements.
The greatest difficulty has always been to devise an instrument
which would accurately record the total change in magnetic flux in
the specimen. The ideal instrument would be a fltixmeter with no
restoring force and no friction. Fluxmeters are on the market in
which the restoring force is negligible only over short periods of time
or in which there is no restoring force but where the friction is appreci-
able; but if it is required that the magnetic cycle have a period of more
than a few seconds, such fluxmeters are out of the question. In
addition they require that the search coil be of such low resistance
that it must ha\-e too few turns for use with long thin specimens, in
which the flux is small. These difliculties have been o\ercome in the
apparatus described below, in which the principal feature is the use of a
'J. A. Ewing, "Magnetic Induction in Iron and Other Metals," 3d ed., p. 118.
2 J. A. Fleming, Proc. Pliys. Soc. Lon., 27, 316-27 (1915).
2 J. B. Johnson, Bell System Tech. Jour., 8, 286-308 (1929).
20
A MAGNETIC CrRVli TRACIili
21
fluxmeter in which the suspended coil has its restoring torque counter-
balanced for all deflections within a range sufficient for accurate
delineation of magnetic curves.
Description of the Apparatus
The operation of the apparatus is as follows: a long, sensitive, photo-
electric cell is fitted with a V'-shaped slit, as shown in Fig. 1 ; a beam of
LAMP
/
FLUXMETER
i— &-
EMF
Fig. 1 — The photoelectric cell circuit.
light is reflected from the mirror of the fluxmeter and focused on the
slit of the photo-electric cell, which is connected, in series with a
source of e.m.f., across the terminals of the fluxmeter; the e.m.f. is
adjusted once for all to such a value that, if the beam is at rest when
at the narrow end of the slit, at any other position the current con-
trolled by the cell will develop a torque in the fluxmeter coil which just
balances the restoring torque of the suspension. The fluxmeter
deflection will then be proportional to the change of flux which has
occurred within the search coil S. It may be found necessary to shape
the slit empirically to correspond to the unequal sensitivities of the
photo-electric cell at different spots. The fluxmeter used is a Leeds
and Northrup type 2290 HS galvanometer. It has a critical damping
resistance of about 100,000 ohms, and when used with about one
hundred ohms in the external circuit it is much over damped.
The apparatus for registering the deflections photographically, and
22
BELL SYSTEM TECIIXICAL JOURNAL
for changing ihe magnetic lield in the specimen, is sIkjwii in I^'ig. 2. A
drum D, carrying photographic paper, is placed in a Hght-tight box
provided with a long, narrow slit parallel to the axis of rotation of the
drum. A beam of light from a second lamp is reflected by the fluxmeter
mirror and focused on the slit. This beam is reflected by the same
mirror which reflects the beam onto the photo-electric cell, the two
X
Fig. 2 — The field current circuit and the piiotographic drum.
beams being incident at different angles. Attached to the shaft of
the drum is an arm A, which slides along the rheostat R. A battery B
is connected across R, and a center tap soldered to it. Between the
arm A and the center tap a varying e.m.f. is produced which is applied
to the held coil F. This e.m.f. reverses its sign e\ery time the arm A
slides past the center of the rheostat, and the latter is curved in a
manner calculated so that the held current will be jjroportional to the
angle of rotation of the drum from the position for zero current. The
.search coil 6" of Fig. 1 is placed within F, and consequently when D is
rotated it moves the photographic paper past the slit so that the
distance moved is proportional to the change in held current, while at
the same time the fluxmeter deflects the beam of light along the slit so
that the deflection is proportional to the time integral of the changes
of flux within S. As the drum is turned from one position to another,
a curve with rectangular axes is thus registered, the scales of which
may be calibrated in terms of B and //. Figs. 4 to 7 are some examples
of curves taken with the apparatus.
In Fig. ?) the electrical circuits are shown in detail. R^ is the rheostat
controlling the held current, and A is the arm which rotates with the
A MAGNETIC CURVE TRACER
23
drum. The battery B-i supplies the field current, and Bz furnishes the
e.m.f. for the photo-electric cell, the value of the potential applied to
the latter being regulated by Ru The potential divider R3, and dry
cell Bu in series with the 10 megohm resistance i?o, are used to balance
out thermo-electric potentials and current from the photo-electric cell
R6
0- 10,000^^
â– AWc^W
0-10,000*^
AMMETER
Fig. 3 — Detailed diagram of the electrical connections.
due to stray light. Ri and R-, are adjusted according to the amount of
flux in the specimen, in order to keep the maximum deflection within
the desired limits. The mutual inductance M is used to balance out
the potentials produced in S when no specimen is within it, so that the
7750
Fig. 4— Hysteresis loop of annealed iron.
24
BELL SYSTEM TECHNICAL JOURNAL
Huxmeter dcHection is proportional to the change in B — II. The
drum is conveniently rotated by an electric motor, connected by geans
so that the drum makes about one revolution in two minutes, and it is
desirable to have this rate variable. The motor may be reversed, so
that complete hysteresis loojis may be recorded.
14,800
-14,800
Fig. 5 — Hysteresis loop of hard iron.
In setting up the apparatus the photo-electric cell may be con-
veniently placed above or below the drum, and one lamp above the
other. The lamp used to illuminate the photo-electric cell should
furnish a brilliant beam, and it was found that a 250 watt Mazda
projection lamp was quite satisfactory.
Fig. 6 — Hysteresis loops of hard iron, with increasing maxinmni fields.
Calibration of the Circuit
The circuit is calibrated by passing a known current through the
primary of a known mutual inductance, the secondary of which is
connected in series with the search coil S. By measurement of the
A MAGNETIC CURVE TRACER
25
deflection produced the rekition between the quantity of electricity
passing through the fluxmeter and its deflection can he determined.
From this relation and other known constants the change in induction
of a magnetic specimen producing a given deflection ma\' he calculated.
This calibration may be done in the following manner: Let the
Fig. 7 — Hysteresis loop of pcrniinvar, sliowing the "wasp waisted" loop.
magnetic specimen be remo\ed, Ri and Ri-, be set on infinite resistance,
the magnetizing coil F be shorted, and a change in the held current
made which will give a convenient deflection on the drum, as shown in
Fig. 8.
Fig. 8 — Line taken for calibrating the apparatus.
Let Im = instantaneous primary current,
to = instantaneous secondary current,
^2 = resistance of secondary circuit,
AI = mutual inductance of M,
L2 = self inductance of secondary circuit.
26 BELL SYSTEM TECIIXICAL JOVRXAL
Then :
Lo dii , M di^f , .
r-2. dt ro dt
Integrating from time / = to / = /„, the time at any later instant,
— I dii -i I diM = — I udt.
''s Jo ''- X Jo '
Now if iM is changed slowly enough
— I dio
'" Jo
is negligible and we have:
- I diM = - 1.(11,
- Jo ^M)
r
or
— IM - - Q^f,
where Qm is the quantity of electricity that has passed through the
fluxmeter in time /q. Now let Q.\f = — K8m, where 5i/ is the deflection
produced when Qm flows. Then:
and
— l,\r — A6.V,
A- = "â– '"
ri^M '
and the quantity of electricity which has passed through the fluxmeter
for any other deflection is
0=-^'s. (1)
'^20.1/
This equation makes it possible to determine B — II, calculated from
Q as described below, by observing the deflection h. Relation (1) may
be determined once for all as it is a constant of the fluxmeter onh-.
The parts of Q passing through R2 and the photo-electric cell will be
negligible on account of their high resistances.
Now suppose a magnetic curve recorded with R(, adjusted until the
deflection is due solely to the magnetization of the specimen. Let the
resistance of the fluxmeter plus that of the secondary of M be denoted
by Rg, and that of S plus R^ be denoted by R,. Then if the field
.1 MAGNETIC CL'RVE TRACI-.R
27
current in is varied slowly enough, the lime lag in the secondary
circuit will be negligible and we shall have for the instantaneous
current in the fluxmeter:
/„ =
Rs + Ra +
R,
Now the e.m.f. in the search coil is
e = - /lA
dt
where A is the cross sectional area of the specimen and N is the number
of turns in the search coil. Then
AN
In =
d{B - //;
dt
and therefore
Q - io
Jo
But by Eq. (1)
therefore
.dl =
R, + Rjl-h^^
- AN
Rs + Rg
Q = - K8
A{B - H) =
\r.
Kb\Rs + RA^ +^
AN
(2)
where
A' =
ro being the total secondary resistance when K was determined. This
equation, then, gives B — II for any given deflection 5, in terms of
known constants. For any fluxmeter, K is determined once for all by
passing the current im through a mutual inductance and measuring the
deflection Bm on a photographic record. The other constants are
changed in a calculable way when the number of turns in the search
coil, the resistance settings, and the cross-sectional area of the sample
are changed.
Sources of Error
Since it is the voltage applied to the magnetizing coil F which is
proportional to the angle through which the drum has rotated, there
28
BELL SYSTEM TECHNICAL JOURNAL
is a lag ill the tield current behind the lield registered on the drum, due
to the self inductance of the coils. Added to this there is a lag in the
secondary due to its self inductance, and another lag due to the time
required for the fluxmeter to act. The effect of these is to widen the
loop. In Fig. 9 is shown a curve traced with no magnetic sample in
the field coil, and with dlljdt so great that the lag is appreciable.
Fig. 9 — Loop made with an air core mutual inductance at a very high d Il/dt.
Fig. 10 shows two loops, the outer one representing a loop as taken on
the apparatus, and the inner one the true loop corresponding thereto.
Let B be some induction near zero, on the traced loop. B will be
incorrect for the indicated value of // by an amount Bo — B, such
that if the field were held constant at that point while the drum con-
tinued to rotate the cur\e would approach Bo as an asymptote, as
indicated by the dotted curve. If dll/dl is not zero, B may be regarded
as momentarily approaching Bo as an asymptote. The equation for
B at any instant is:
\,~ + B = Bo, (3)
where Xj is the time constant of the circuit and Bo is not a constant
but a function of // and /. If we assume that dB/dll is constant for a
small region in the neighborhood oi B = Q, we have, putting Allc
equal to the error in coercive force lie,
Bo- B =^j-^JJc
.1 MAGXETIC CURVE TRACIili
Combining this with Eq. (3), we have
A//, = Xi
(IH
dt
29
(4)
Data taken with no magnetic specimen inserted show that this Hnear
relation actually exists. Added to this there is an increase in II ^ due to
Fig. 10 — A diagram to illustrate the widening of a loop due to inductance.
eddy current lag. Johnson ^ has derived an equation for this, and with
a slight modification to make it applicable to cylindrical specimens, it
is:
nio-3 ,dBdH
2 p dtl di
(5)
where p is the resistivity of the specimen, and r its radius. This gives
us for the total error.
Mh = Xi +
n 10-9 ^dB\dII
2 <.â– ' dllj dt
= (Xi + X.)
clU
dt
This equation was tested experimentally by taking a series of loops
30
BELL SYSTEM TECIIXICAL JOURNAL
with varying dH/dl. The specimen used was a cyhnder of 81 per cent
Ni permalloy, 60 cm. long and 0.1 cm. in diameter, and was placed in a
magnetic yolk. Its hysteresis loop, as shown in Fig. 11, has an
8700
B
-0.2
I'ig. 11 — A hysteresis loop of permalloy containing 81 per cent nickel.
unusual slope, 225,000 at 5 = 0. This gives X2 = .055 sec. From
this series of curves the straight line shown in Fig. 12 was obtained, for
which Xj + X> = 0.314 sec. By another set of loops in which the
08
01
.02
05
.03 .04
dH/dt
Fig. 12 — The change in apparent lie with varying dlljdl
.06
.07
A MAGNETIC CURVE TRACER
31
deflection is produced by an air core mutual inductance, Xi is found to
be 0.134 sec. This determines Xo as 0.180 sec, in disagreement with
the value 0.055 sec. calculated from Eq. 5. Johnson assumes in his
derivation that dB/dll is constant and hence that the shape of the
curve before He is reached has no effect on A/Zc- It is probable that
if the equation were changed to allow for dB/dll being a function //,
10,000
8,000
^
.^
""^
^
6,000
f
4,000
'
2,000
B
1
1
-2,000
-4,000
/
\
-6,000
,.â– '
^^
-8,000
-10,000
jgnffiSC-
-0 5-0 4 -0.3 -0.2 -0.1 0.1 0.2 0.3 0.4 0.5
H
Fig. 13— A hysteresis loop of permalloy containing 78.1 per cent nickel.
that the difference could be accounted for. At any rate, this error is
negligible for all but specimens with exceptionally high dBidH or
great thickness, and the true coercive force can always be found by
taking two loops with different values of dHldt and extrapolating to
dllldt = 0.
Another possible source of error is the passage of a large fraction of
the photo-electric cell current through the search coil, the field being
thereby altered. The maximum photo-electric cell current used is on
the order of 5(10)-^ amperes. Since the search coil is unlikely to have
more than about 400 turns per centimeter, this would make the
maximum error in // about 2.5(10)-* gauss, which is negligible for most
measurements.
As a test of the accuracy of the instrument, a comparison was made
with curves made by ballistic galvanometer measurements. Fig. 13
shows a loop taken of the specimen which Bozorth used in some
previous measurements.-* ^ Both the coercive force and the maximum
induction taken by the two methods agreed to within less than one
' R. M. Bozorth, Phys. Rev., 32, 124-132 (1928).
32
BELL SYSTEM TECHNICAL JOURNAL
per cent. Fij^. 14 shows an initial magnetization curve wliich gives a
value of the initial permeability agreeing accurately with the value
determined ballistically.
A Huxmeter with no restoring torque is also useful in certain t>'pes of
current measurements. If the average value of a current which
fluctuates too nuich to he read on a slowly moving meter is desired, it
6000
5000
4 000
B 3000
2000
1000
0.5
Fig. 14 — All initial magnetization curve of tlie specimen of 78.1 per cent
nickel permalloy.
can be integrated on the fluxmeter, and the average value obtained by
dividing the total quantity of electricity which has passed through by
the time during which the measurement was made. Also if a current is
too small to be read directly on a galvanometer it may be possible to
maintain it for a sufficient length of time to give a readable deflection
on the fluxmeter, and again the current will be obtained by dividing
by the time.
In conclusion I wish to thank Dr. R. M. Bozorth for suggestions
given during the development of the apparatus, and Mr. A. W. Metz
for his assistance in taking the curves.
A Multi-Channel Television Apparatus *
By HERBERT E. IVES
A bar to the attainment of television images having a large amount of
detail is set by the practical difficulty of generating and transmitting wide
frequency bands. An alternative to a single wide frequency band is to
divide it among several narrow bands, separately transmitted. A three-
channel apparatus has been constructed in which prisms placed oyer the
holes in a scanning disc direct the incident light into three photoelectric cells.
The three sets of signals are transmitted over three channels to a triple elec-
trode neon lamp placed behind a viewing disc also provided with prisnis
over its apertures so that each electrode is visible only through every third
aperture. An image of 13,000 elements is thus produced. For the suc-
cessful operation of the multi-channel system, it is imperative to have very
accurate matching of the characteristics in the several channels.
IF, in a received television image, the individual image elements are,
as they should be, of such a size as to be just indistinguishable, or
unresolved, at a given observing distance, the number of image ele-
ments increases directly with the area of the image. The number of
such indistinguishable elements in everyday scenes, in the news
photograph, or in the frame of an ordinary motion picture is aston-
ishingly large. An electrically transmitted photograph 5 inches by 7
inches in size, having 100 scanning strips per inch, has a held of view
and a degree of detinition of detail, which, experience shows, are
adequate (although with little margin) for the majority of news event
pictures. It is undoubtedly a picture of this sort that the television
enthusiast has in the back of his mind when he predicts carrying the
stage and the motion picture screen into the home over electrical
communication channels. In this picture, the number of image
elements is 350,000. At a repetition speed of 20 per second (24 per
second has now become standard with sound films) this means the
transmission of television signals at the rate of 7,000,000 per second,—
a frequency band of Syi million cycles on a single sideband basis.
This may be compared to the 5,000 cycles in each sideband of the
sound radio program, or it may be evaluated economically as the
equivalent of a thousand telephone channels.
When we examine what has been achieved thus far in television, we
find that the type of image successfully transmitted falls very far
short of the finely detailed picture just considered. Probably the
most satisfactory example of television thus far demonstrated is the
* Jour. Optical Soc, Jan., 1931.
33
34 BELL SYSTEM TECHNICAL JOURNAL
72-line picture used in the two-way television-telephone installation of
the American Telephone and Telegraph Company in New York.*
Here the object to be transmitted is definitely restricted to the human
face, which tills the whole field of view, and is adetjuately rendered by
the 4,sS00 image elements used.
The gap between the 4.000 elements of this image and the 350,000
considered abo^â– e is enormous, not only in figures, but in terms of
technical j)ossil)ilit\' of bridging. I^\cn if we are forced to content
ourscUes with relatively simple t>i)es of scenes for television trans-
mission, still the fact must be s(]uarely faced that a very much larger
numbci' of image elements nuist be transmitted than h.is thus far been
found possible; and a far wider frequenc\' band utilized than has e\er
been used in any communication problem. Now the situation is,
simply stated, that all parts of the television system arc already having
serious difficulty in handling the 4,500-element image. Consequently,
a major problem in television progress is to develop means to extend the
practical frequency range.
It will be worth while to survey briefly the points in a television
system where ditiiculty is now encountered when the attempt is made
to increase the amount of image detail and the accompanying band of
transmitted frequencies. Consider in turn the scanning discs at
sending and receiving ends, the photoelectric cells, the amplifying
systems, the transmission channels, the receix'ing lamps.
In the scanning disc at the sending end, which we shall assume
arranged for direct scanning, increased detail means either loss of
light or increase in the size of the disc. In either case, the factor of
change involved is large. For instance, if the number of scanning
holes is doubled in a disc of given size, providing four times the number
of image elements, the holes must be spaced at half the angular distance
apart, and twice the number of holes, imagined placed end to end,
must be included in this half diameter scanning field. The holes will
therefore be of one-quarter the diameter or 1/16 the area. The light
falling on the photoelectric cell at any instant is the light transmitted
by one hole; in this case, 1/16 the amount with the disc of half the
number of holes. In general, the light transmitted by the disc to the
cell decreases as the square of the number of image elements. If the
disc is enlarged so as to hold the transmitted light unchanged, its
diameter increases directly as the number of image elements. It is
obvious that any considerable increase in the number of image ele-
ments — such as ten times — demands either enormously increased
sensiti\'eness in our photo-responsive de\'ices, or cjuite fabulous sizes of
I Bell System Techuieal Jounuil, July !<),>(), p. 448.
A AIULTI-CIIANNEL TELEVISION APPARATUS 35
discs. Perhaps the most pertinent conckision from this survey is that
the disc, while ciuite the simplest means for scanning images of few
elements, is entirely impractical when really large numbers of image
elements are in question. As yet, however, no practical substitute
for the disc of essentially different character has appeared.
Turning now to the photoelectric cell. The question of adequate
sensitiveness to handle a large number of image elements is intimately
connected with the method of scanning, as has just been brought out,
so that no simple answer is possible. It is, however, probable that a
very considerable increase in sensitiveness over anything now available
must be anticipated, whatever scanning device is adopted. In the
matter of frequency range there is definite information.- In cells
depending on gas amplification (such as argon or neon) a characteristic
behavior is a falling off of output with frequency, greater the higher
the voltage used, which, becoming noticeable at about 20,000 cycles,
may at 100,000 cycles be so considerable as to constitute a practical
block to transmission. Vacuum cells are free from this failing, but
are much less sensitive. Systematic experiment and development of
photoelectric cells with particular reference to extending their range of
frequency response is indicated as a necessary step in the attainment
of a many-element image.
Taking up next the circuits associated with the photoelectric cell, we
find, in general, that the higher frequencies progressively suffer from
the electrical capacity of cells and associated wiring and amplifier
tubes. This in turn calls for auxiliary equalizing circuits, with
attendant problems of phase adjustment, and for increased amplifica-
tion. Amplifiers capable of handling frequency bands extending from
low frequencies up to 100,000 cycles or over offer serious problems.
Communication channels, either wire or radio, are characterized by
increasing difficulty of transmission as the frequency band is widened.
In radio, fading, different at different frequencies, and various forms of
interference stand in the way of securing a wide frequency channel of
uniform efficiency. In wire, progressive attenuation at higher fre-
quencies, shift of phase, and cross-induction between circuits offer
serious obstacles. Transformers and intermediate amplifiers or re-
peaters capable of handling the wide frequency bands here in question
also present serious problems.
At the receiving end of the television system, conditions are similar
to the sending end. The neon glow lamp, commonly used for re-
ception, is already failing to follow the television signals well below
40,000 cycles, and, in the case of the 4,500-element image above
2 Loc. cit., p. 456.
36 BEI.L SYSTEM TECHXICAL JOURXAL
referred to, the neon must be assisted by a frequently renewed ad-
mixture of hydrogen, which again cannot be expected to increase the
frequency range indefinitely. In the scanning disc, as at the sending
end, increasing the number of image elements rapidly reduces the
amount of light in the image. With a plate glow lamp of given
brightness, the apparent brightness of the image is inversely as the
number of image elements.
From this rapid survey, it is clear that at practically every stage in
the television system, we encounter serious difficulties when a large
increase in image elements is contemplated. It is not claimed that
these difficulties are insuperable. One of the chief uses of a tabulation
of difficulties is to aid in marshalling the attack upon them. But the
existing situation is that if a many-element television image is called
for today, it is not available, and one of the chief obstacles is the difficulty
of geiterating, transmitting, and recoverijig signals extending over wide
frequejicy hands.
One alternative, which prompted the experimental work to be
described below, is the use of multiple scanning, and multiple-channel
transmission. The general idea, which is obvious from the name given
to the method, is to divide the image into groups of elements, the
various groups to be simultaneously scanned, and to transmit the
signals from the several groups through separate transmission channels.
In place of apparatus to generate and transmit a frequency band of n
cycles, we arrange m scanning processes each to provide frequency
bands of njm cycles width ; njm being chosen as within the limits set by
the available practical elements of a television system. It will appear
that the method which has been developed does provide an image of
manyfold more image elements than heretofore, and may make easier
the problem of transmission over practical transmission lines.
Description of a Three-Channel Apparatus
The multi-scanning apparatus which has been constructed and
given experimental test uses scanning discs over whose holes are
placed prisms of several different angles. At the sending end, the
beams of light from successive holes are thereby diverted to different
photoelectric cells. At the receiving end, the prisms similarly take
beams of light from several lamps and divert them to a common
direction. The mode of action of the prisms is illustrated in Fig. la,
where a three-channel arrangement is shown, which is that actually
used in the experimental apparatus. In the figure, the disc holes are
shown disposed in a spiral, at such angular distances apart that
three holes are always included in the frame/. Over the first hole of a
A MULT I- CHANNEL TELEVISION APPARATUS 37
set of three is placed a prism Fi which diverts the normally incident
light upward; the second hole is left clear; the third is covered by a
prism P2 turned to divert the light downward. If wc imagine the
prisms removed and a single channel used instead of the three that are
proposed, it is clear that the holes would have to be spaced three times
as far apart so that no more than one would be included in the frame/
at one time. The diameters of the holes, and the radial separation of
the first and last in the spiral would be unchanged. Quite apart,
therefore, from the smaller frequency bands which are sufficient to
carry each of the three sets of signals, which is the principal objective
sought, there is realized in this arrangement a reduced size of apparatus
for the same size of disc holes.
Studying more closely the division of the light into three sets of
beams, it is important to note that the signals transmitted by any one
of the three sets of holes are continuous — as one hole of a given prism
series passes out of the frame the next of the same series comes in.
The signals generated in each photoelectric cell are accordingly exactly
like those of a single-channel system.
Before describing the details of the apparatus, the general relation-
ship between the number of image elements, band width, number of
channels, and shape of picture may be developed. For this purpose,
let the following symbols be used.
B = frequency band available in one channel, in cycles per second.
F = repetition frequency of images, per second.
C = number of communication channels.
n = total number of scanning holes.
ajb = ratio of tangential to radial dimensions of frame.
a — angular opening of frame.
We shall assume that the picture elements into which the frame is
imagined divided are symmetrical in shape, i.e. either circles or squares.
We then have that
the number of picture elements in the radial direction = number of
holes = n\
the number of picture elements in the tangential direction = {alb)-n.
Now the number of signal cycles corresponding to this number of
elements is (1/2) • {a[b)n.
The number of cycles per second in one transit along the frame
= {\l2)-{alb)-n'F-
over the whole picture it is (1/2)' (ajb)-n- F-n = {l/2)ia/b)Fn-;
38
BKLL SYSTEM TECUM CAL JOURNAL
and the nuniljcr (jf c\cles per second for each ciiannel = (1/c)
•(l/2)(a/6j^«2 ^'^
The angular opening of the frame a = 36()/w X C.
Tlie number of picture elements = n~'{alb).
These formuke ma>' he utilized upon assuming values for any of the
variables, to fix the values of the other. In the present case, it was
decided for reasons of simplicity to restrict the number of channels to 3.
L2L1
l-c
1-b
Fig. 1 — Schematic of three-channel television apparatus, {a) Receiving end
disc with spiral of holes provided with prisms. (6) Sending end disc with circle of
holes provided with prisms, (c) General arrangement of apparatus.
The band width was chosen as that found feasible in the two-way
tele\'ision system, namely 40,000 cycles. The picture shape chosen
was that of the sound motion picture, for which ajb = 1 j6. The
repetition frequency assumed was 18 per second, again following
closely that of existing experimental synchronizing apparatus. Sub-
stituting these w'llues in the formula rearranged to give ;/, we get for
the nimiber of holes,
and for a,
ilBbc
F
^a
= 108
— X 3 = 10 degrees,
for the number of picture elements.
n = (108)2 X ^ = 13,608.
In utilizing the prism disc principle at the sending end, direct
A MULTI-CHANNEL TELEVISION APPARATUS 39
scanning, in which the object is imaged on the disc, was chosen, since
beam scanning would introduce the problem of separating the light
reflected from the object from the several spots simultaneously pro-
jected from the disc. Since the light going through the disc must be
separated into several beams to be directed into separate photoelectric
cells, the full aperture of the image forming lens must be di\ided by C,
the number of channels, with a consequent proportional loss of light to
each cell. (This loss counterbalances the decreased size of disc above
noted.) It therefore becomes necessary to insure a very high illumi-
nation of the object. In the present case, it was decided to use motion
picture him to provide the sending end image, since this can have a
large amount of light concentrated through it by an appropriate lens
system.
The use of motion picture him permitted a simplification of the
transmitting disc, which is illustrated in Fig. lb. This consists in
arranging the scanning holes in a circle instead of a spiral, and pro-
ducing the longitudinal scanning of the film by giving it a continuous
uniform motion at right angles to the motion of the scanning holes.
The continuous motion of the film also avoids the loss of transmission
time that an intermittent motion demands for the shutter interval.
At the receiving end, a spiral of holes is used as shown in Fig. la.
There again, because of the division of the light into three beams, the
angle which can be subtended by the light source (neon lamp) is much
restricted. In consequence, the neon lamp cathodes are of small area,
and a lens system has been used to focus their images on the pupil of
the observer's eye. Other methods of receiving, which promise to be
less restricted as to position of observation, are possible, however, as
discussed below.
With this surve\' of certain of the more important features of the
system, we may proceed to a more detailed account of the apparatus as
constructed. The general arrangement of parts is shown in Fig. Ic
and in the photographs, Figs. 2, 3, 4 and 5 in all of which the symbols
are uniform. Both sending and receiving discs were, for simplicity of
operation, mounted on the same axis, driven by the motor M. This
means that no question of synchronization entered. Synchronization
is in fact a separate problem, having nothing to do with multi-channel
operation and has been very completely solved in connection with other
television projects.^ If it should be decided to transmit the multi-
channel image to a distant point, the apparatus could be cut in two
and each end, after separation to the desired distance, operated by
synchronous motors controlled in approved fashion. Similarh', no
long transmission lines were included.
40
BELL SYSTEM TECHNICAL JOURNAL
Starting at the extreme right end of the schematic drawing Fig. \c,
we have an arc lamp A, a cyHndrical lens Li, a condensing lens L2, the
two lenses together concentrating a line of light on the film F. Be-
tween the film and the disc is a lens Ls which images the film on the
disc. Directly behind the disc Di, with its circle of prism covered
holes, is a second cylindrical lens L^ which concentrates the transmitted
Fig. 2 — Sending end of three-channel television apparatus, showing film driving
arrangements.
light laterally, upon the three photoelectric cells ^i, 52, Sz. By virtue
of this lens arrangement, the light falls upon the cells in three small
practically stationary spots. Additional apparatus not shown in the
diagram but visible in the photographs are gears by means of which the
film is driven from the disc axle through a differential, which permits
the film to be framed up and down. The light beam is directed through
the film at right angles to the axis of the discs by means of two prisms,
w^hereby certain conveniences in driving and handling the film are
attained.
The photoelectric cells are similar to ones previously described.
The amplifier system was substantially identical with that used in the
two-way television system, and need not be described again. Simi-
A MULTI-CHANNEL TELEVISION APPAR.ATUS 41
larly, the amplifiers at the receiving end were the actual set used in the
three-color television apparatus previously described.^
At the receiving end, the three sets of signals were supplied to the
three electrodes of a special neon lamp N, shown in Fig. 5, which is
pro\-ided with a hydrogen valve to enable it to respond to the higher
frequencies. Condensing lenses L5 and Lo image the three electrodes
Fig. 3— Sending end of three-channel television apparatus, showing sending prism
disc and photoelectric cells.
at the eye, where another lens L- is placed at the eye to focus the face
of the disc D^. By this system, nine electrode images are formed, of
which three are superposed at the eye, and successive scanning holes are
seen illuminated by each electrode in turn. This viewing arrangement,
by which the image is visible to only a single eye, is adequate for an
experimental investigation of the multi-channel method, but some other
scheme would of course be needed if the method were developed into a
practical form. Of several schemes, mention will be made here only
5 Journal oj the Optical Society, February, 1930, p. 11.
42
BELL SYSTEM TECHNICAL JOURNAL
of the possible use of a triple grid of neon tubes, using a triple distrib-
utor of the type used in displaying images to a large audience in our
initial work in 1027. •*
Discussion of Results
The three-channel apparatus, when all parts are properly function-
ing, yields results strictly in agreement with the theory underlying
its construction. The 13,500-element image, in resolving power and
Fig. 4 — Receiving end of three-channel tele\ision apparatus.
amount of detail handled, is a marked advance over the single-channel
4,500-element image. Even so, the experience of running through a
collection of motion picture films of all types is disappointing, in that
the number of subjects rendered adequately by even this number of
image elements is small. "Close-ups" and scenes showing a great
deal of action, are reproduced with considerable satisfaction, but
scenes containing a number of full length figures, where the nature of
the story is such that facial expressions should be watched, are very
* Bell System Technical Journal, October, \^11 , pp. 551-652.
A MULTI-CHAyXEL TELEVISION APPARATUS
43
Fig. 5 — Three-electrode neon lamp used for three-channel television reception.
44 BKI.L SYSTEM TRCIINICAL JOi'RXAL
far from satisfactory. On the whole, the general opinion expressed in
an earlier paragraph is borne out, that an enormously greater number
of elements is required for a television image for general news or
entertainment purposes. This, how'ever, was anticipated, and the
real question is whether the results of this experiment indicate that
the finer grain image is best attained by resort to multi-channel means.
This leads to a discussion of what has proved to be a serious practical
difficulty with the multi-channel apparatus. This is the problem of
keeping the several channels properly related to each other in signal
strength. In the e.xperimental apparatus, the direct current com-
ponents (introduced at the receiving end) and the alternating current
signals, are separately controlled, manually, by potentiometers.
These have fine enough steps so that with care, with a non-changing
image, a uniform picture may be obtained. If, however, for any
reason the signals on one of the channels becomes too strong or too
weak, the picture exhibits at once a strongly lined appearance. The
eye is quite sensitive to irregularity of this sort, and the transition
from a smooth grainless image to one showing a periodicity of 1/3 the
number of constituent lines largely offsets the higher resolving power
afiforded by the actual number of scanning lines used. A characteristic
practical defect of the system as set up is that any marked change in the
general character of the signal, such as is produced by a shift from
close-up to a wide angle view may throw out the existing signal
balance sufficiently to show objectionable grain in the picture.
DifTerences of this sort in the three signals are of course caused in
general by differences in the characteristics of the three circuits. Such
differences can arise from overloading of amplifier tubes, whereby one
or more may be working on a non-linear portion ; by rectifying action
of different amounts in the tubes immediately associated with the neon
lamps, or in the neon lamp electrodes themselves. A remedy is the
careful design and test of all parts of the system to insure the greatest
possible uniformity of performance. When this is carefully done, the
behavior of the three signals is reasonably satisfactory.
Conclusion
We are, as a consequence of this work, in a position to make a
general comparison of the two chief theoretical means for achieving a
television image of extreme fineness of grain, which are (1) extension
of the frequency band, and (2) the use of several relatively narrow
frequency bands. Both, because of the diminished amount of light
which finer image structure entails, demand enhanced sensitiveness of
the photo-sensitive elements at the sending end, and increased efficiency
A MULTI-CHANNEL TELEVISION APPARATUS 45
fo the light sources at the receiving end. The multi-channel scheme
described has some advantage in compactness over the equivalent
single-channel apparatus, but since it is restricted to narrow angles of
illumination of the discs the overall efficiency of light utilization is not
essentially different. Comparing now the demands made upon the
electrical systems the differences between the two methods are clear
cut. Method (1) demands an extension of the frequency range of all
parts of the apparatus, the attainment of which depends upon physical
properties and technical devices whose mastery lies in the indefinite
future. Method (2) demands a multiplication of apparatus parts, and
careful design and construction of these parts so as to insure accurately
similar operation of a considerable number of electrical circuits and
terminal elements. The attainment of the necessary uniformity of
performance of the several electrical circuits and terminal elements,
while involving no fundamental problems, must present increasing
difficulty with the number of channels used.
Condenser and Carbon Microphones — ^Their Construction
and Use *
By W. C. JONES
Of the numerous microphones which have been developed since Bell's
original work on the telephone, only two are used extensively in sound
recording for motion pictures, namely, the condenser microphone and the
carbon microphone.
The condenser microphone was first proposed in 1881 but owing to its
low sensitivity was limited in its field of usefulness until the development
of suitable amplifiers. In 1917, K. C". W'ente published an account of the
work which he had done on a condenser microphone having a stretched
diaiihragm and a back plate so designed as to introduce an appreciable
amount of air damping. The major portion of the condenser microphones
used today in sound recording embody the essential features of the Wente
microphone. Marked progress has, however, been made in the design and
construction of these instruments with the result that they are not only more
sensitive but also more stal)le. The factors which contribute to this im-
provement are described in detail in this paper. Recently a number of
articles ha\e appeared in the technical press calling attention to certain
discrepancies between the conditions under which the thermophone calibra-
tion of the condenser microphone is made and those which exist in the studio.
The nature of these discrepancies and their bearing on the use of the micro-
jjhone are discussed.
Microphones in which the sound pressure on the diaphragm produces
changes in the electrical resistance of a mass of carbon granules interposed
between two electrode surfaces have been used commercially since the
early days of the telephone. In recent years the faithfulness of the repro-
duction obtained with the carbon microphone has been materially improved
by the introduction of an air damped, stretched diaphragm and a push-pull
arrangement of two carbon elements. This instrument is finding extensive
use in sound recording and reproduction fields where carbon noise is not an
important factor. The outstanding design features of the push-pull carbon
microphone are described in this paper and suggestions made as to the
precautions to be taken in its use if the best (luality, maximum life, etc.
are to be obtained.
OF the numerous microphones which have been developed since
Bell's original work on the telephone, only two are used exten-
sively in sound recording for motion pictures, namely, the condenser
microphone and the carbon microphone. It has therefore been
suggested that it would be fitting to review at this time the con-
struction of these instruments and consider some of their trans-
mission characteristics and the precautions which should be exercised
in their use.
Condenser Microphone
In 1881, A. E. Dolbear ' proposed a telephone instrument which
could be used either as an electrostatic microphone or receiver. This
* Presented at Soc. of Motion Picture Engineers' Convention, Oct. 20, 1930;
Journal, Soc. of Motion Picture Engineers, Jan., 1931.
^ "A New System of Telephony," A. E. Dolbear, Scientific American, June 18,
1881, p. 388.
46
CONDENSER AND CARBON MICROPHONES 47
instrument consisted of two plates insulated from one another and
clamped together at the periphery. The back plate was held in a
fixed position whereas the front was free to vibrate and served as a
diaphragm. It is obvious that, if the diaphragm were set in vibration
by sound pressure, the electrical capacitance between the two plates
would be changed in response to the sound waves, and if a source of
electrical potential were connected in series with the instrument a
charging current would fiow which would be a fairly faithful copy ot
the pressure due to the sound wave. Apparently Dolbear realized
that the current developed in this way would be minute, for in the
telephone system which he proposed as a substitute for the one using
Bell's magnetic instruments he employed the electrostatic instrument
only as a receiver and adopted the loose contact type of microphone.
At approximately the same time an article appeared in the French
press - calling attention to the use of a condenser as a microphone and
commenting on the fact that this type of microphone had been found
to be less sensitive than the loose contact type.
Owing to the low sensitivity of the condenser microphone, the field
of usefulness of this instrument was extremely limited for a number
of years and it did not assume a position of importance among the
instruments used in acoustic measurements and sound reproduction
until suitable amplifiers had been developed. The development of
the vacuum tube amplifier, however, filled this need. In 1917 E. C.
Wente ^ published an account of the work which he had done on an
improved condenser microphone having a stretched diaphragm and a
back plate so located relative to the diaphragm that in addition to
serving as one plate of the condenser it added sufficient air damping
to reduce the eft'ect of diaphragm resonance to a minimum.* The
response of this instrument was sufficiently uniform over a wide range
of frequencies to make it not only useful in high quality sound repro-
duction but a valuable tool in acoustic measurements in general.
The major portion of the condenser microphones used today in
sound recording embody the essential features of the Wente micro-
phone. Marked progress has, however, been made in the design and
construction of these instruments since the initial disclosure and it
will no doubt be of interest to many to consider briefly the nature of
this advance.
2 "La Lumiere Electrique," 1881, p. 286.
3 "A Condenser Transmitter as a Uniformly Sensitive Instrument for the Absolute
Measurement of Sound Intensity," E. C. Wente, Physical Review, July 1917, pp.
39-63. "Electrostatic Transmitter," E. C. Wente, Physical Review, May 1922, pp.
498-503.
■» A discussion of the theory of air damping is given in "Theory ot \ ibratmg
Systems and Sound," 1. B. Crandall, pp. 28-39.
48 BELL SYSTEM TECHNICAL JOURNAL
In the early microphones employing air damping the diaphragm was
composed of a thin sheet of steel which was stretched to give it a
relatively high stiffness. When assembled in the microphone the
stiffness was further increased by that of the air film between diaphragm
and the damping plate with the result that the resonant frequency
was well above the frequencies which it was desired to transmit and
the diaphragm vibrated in its normal mode over a wide frequency
range. In such a structure the mechanical impedance for frequencies
below resonance is due almost entirely to stiffness reactance. Hence a
constant sound pressure produces substantially the same displacement
of the diaphragm at all frequencies within this range and uniform
response results except at the very low frequencies where an appreciable
reduction in the stiffness of the air film occurs. The effective mass of
a steel diaphragm is, however, relatively large and necessitates a
comparatively high stiffness to secure the desired resonant frequency.
From the standpoint of securing maximum sensitivity of the micro-
phone, i.e. displacement of the diaphragm per unit force, it is of course
important to make the stiffness as low as possible and employ as small
a value of mechanical resistance as is consistent with the degree of
damping required. An improvement in both respects can be effected
by decreasing the mass of the diaphragm for with a reduced mass a
given resonant frequency can be obtained with lower values of stiffness
and the desired damping constant secured with less mechanical
resistance.
The aluminum alloys have therefore replaced steel in the diaphragms
of most of the condenser microphones in use today. A typical example
of such a microphone is the Western Electric Company's instrument
(394-type) shown in the photograph. Fig. 1, and the cross-sectional
view, Fig. 2. The diaphragm of this instrument is made from alu-
minum alloy sheet .0011 inch in thickness. The edges are clamped
securely between threaded rings, gaskets of softer aluminum being
provided to prevent damage at the clamping surfaces. The requisite
stiffness is obtained by advancing the stretching ring until a resonant
frequency of 5,000 cycles is obtained. The method of determining
the resonant frequency of the diaphragm is as follows. The diaphragm
assembly to be tested is coupled to a condenser microphone which is
provided with a suitable circuit for measuring its output. A special
telephone receiver is placed in contact with the diaphragm on the
side opposite to the coupler. Current from a vacuum tube oscillator
is then passed through the winding of the receiver, setting up eddy
currents in the diaphragm under test. The forces which are developed
as a result of the reaction of the magnetic field produced by the eddy
CONDENSER AND CARBON MICROPHONES
49
currents and that of the permanent magnet of the receiver set the
test diaphragm in motion. The resonant frequency is determined by-
noting the frequency at which the output from the condenser micro-
phone is a maximum.
In the early Wente microphone the damping plate was a continuous
surface. Subsequent work by I. B. Crandall ^ showed that the re-
quired amount of damping at the resonant frequency could be obtained
without adding unduly to the impedance at other frequencies by cut-
ting grooves in the plate. This reduced the stiffness introduced by the
air film and decreased the irregularity in response at low frequencies
previously mentioned. The grooves in the damping plate of the
Fig. 1 — Western Electric Company's 394-type condenser microphone.
Western Electric Company's 394-type microphone are cut at right
angles. Holes, tapered at the outer end to reduce resonant effects,
are bored through the plate at the intersection of the grooves to form
connecting passages between the air film at the front and the cavity
at the back. In order to prevent the resonance which would result
if the grooves extended into the portion of the chamber surrounding
the damping plate, the outer ends are closed by an annular ring which
is pressed over a shoulder on the plate. The surface of the damping
plate is plane within 8 X 10"^ inch. The departure from a plane in
any individual case is determined commercially by the interference
pattern developed when an optically flat plate is placed over the
damping plate under test.
* "The Air Damped Vibratorv System: Theoretical Calibration of the Conilenser
Transmitter," I. B. Crandall, Physi'cal Rcvinc, June 191S, pp. 449-46U.
50
BELL SYSTEM TECHNICAL JOURNAL
A duralumin spacing ring .001 inch in thickness separates the
damping plate from the diaphragm. It is essential that all dust and
dirt be excluded from this space. To prevent foreign material from
entering through the holes in the plate a piece of silk is fastened over
the outer surface. The asseml)l\' of the diaphragm and the damping
plate is made in a dust-proof glass cabinet.
If the back wall of the condenser microphone were rigid, changes in
the separation between the damping plate and the diaphragm of
sufficient magnitude to affect not only the sensitivity of the instrument
but also its frequency response characteristic would result from vari-
ations in barometric pressure. Complete compensation for these
COMPENSATING
DIAPHRAGM
DIAPHRAGM
DAMPING
PLATE GROOVE
Fig. 2 — Cross-sectional view of the 394-type condenser microphone.
changes in pressure can only be obtained by permitting free inter-
change of air between both sides of the microphone diaphragm.
This is, however, objectionable owing to the fact that sufficient
moisture is likely to be introduced to start corrosion and affect the
insulation between the damping plate and the diaphragm. A com-
pensating diaphragm of organic material has therefore been introduced
which prevents this undesirable effect of humidity but is sufficiently
low in stiffness to equalize the changes in pressure encountered in the
normal use of the microphone.
In order to prevent transmission losses at voice frequencies due to
the presence of the compensating diaphragm, an acoustic valve is
CONDENSER AND CARBON .UICROPIIONES
51
inserted between the damping plate and this diaphragm. This valve
consists of a disc of silk clamped between two aluminum plates of
unequal diameters, (ias in passing from the damping plate to the
compensating diaphragm moves laterally from the edge of the smaller
plate through the silk to a hole in the center of the larger plate. The
impedance of this path is high at voice frequencies but low enough for
steadily applied pressure differences to permit compensation for changes
in barometric pressure.
After the damping plate and diaphragm are assembled the space
between the clamping rings is tilled with beeswax to make the joints
gas-tight and exclude moisture. A hole is, however, provided for
filling the microphone with nitrogen. The purpose of the nitrogen is
to prevent corrosion of the damping plate and diaphragm surfaces
and eliminate any reduction in pressure due to oxidation of the sealing
compound.
It has been customary for some time to determine the response
characteristics of a condenser microphone by the thermophone
method.*^ In making this measurement the diaphragm of the micro-
phone is coupled acoustically to the thermophone in the manner
shown in Fig. 3. The thermophone consists of two strips of gold foil
CONDENSER
TRANSMITTER'
TO VACUUM TUBE
VOLTMETER
THERMOPHONE
OF GOLD FOIL
TO VOLTAGE
SUPPLY FOR
RESISTOR
INLET OUTLET
HYDROGEN
Fig. 3 — Cross-sectional view of the thermophone and the condenser microphone.
which are mounted on a plate and fit into the recess in the front of
the microphone. Capillary tubes are provided for filling the space
enclosed between the plate and the microphone diaphragm wnth
•^ "The Thermophone as a Precision Source of Sound," H. D. Arnold and I. B.
Crandall, Physical Review, July 1917, pp. 22-38. "The Thermophone," E. C.
Wenle, Physical Review, April 1922, pp. 333-345. "Speech and Hearing," H.
Fletcher, 1929, Appendix A.
52 BULL SYSTEM TECHNICAL JOURNAL
hydrogen. Tliis is done in order to make the wave-length of the sound
developed in the recess as large as possible compared with dimensions
of the chamber. If this were not the case the sound pressure at dif-
ferent positions in the chamber would not be in phase and the condi-
tions on which the computations of the magnitude of the sound
pressure are based would not be met. A direct current of known
value is passed through the foil. Superimposed upon the direct current
is an alternating current of the desired frequency which causes fluctu-
ations in the temperature of the foil and in the gas immediately
surrounding it. These tluctuations in temperature in turn cause
changes in the pressure on the microphone diaphragm. The magni-
tude of the pressure developed on the diaphragm can be computed
from the constants of the thermophone and the coupling cavity, and
the voltage developed by the microphone for a given pressure deter-
mined with suitable measuring circuits.'^ Obviously, such a calibration
affords a measure of the response of the microphone in terms of the
actual pressure developed on the diaphragm and is independent of the
external dimensions of the instrument. Hence, it does not take into
account any effect which the microphone may have on the sound field
when used as a pick-up instrument for recording or broadcasting pur-
poses. The thermophone calibration is often referred to as a "pres-
sure" calibration and the response obtained by placing the instrument
in a sound field of constant pressure, a "field " calibration. A thermo-
phone calibration of a representative Western Electric 394-type con-
denser microphone is shown on Fig. 4.
For many of the uses to which the condenser microphone is put, for
example the calibration of head type telephone receivers, the condi-
tions under which it operates agree with those under which the thermo-
phone calibration is made. There are, however, cases where this
agreement does not exist, for when a microphone is inserted in a
sound field of uniform intensity the pressure on the diaphragm may
depart rather widely from a constant value in certain frequency
ranges. Several articles ** have recently appeared calling attention to
this discrepancy between the pressure and field calibrations and
pointing out that a pressure calibration of a microphone may not be
entirely representative of its performance under the conditions which
exist in a studio.
'• "Master Reference System for Telephone Transmission," W. H. Martin and
C. H. G. Gray, Bell System Technical Journal, July 1929, pp. 556-559.
"^''The Use of a VVente Condenser Transmitter to JNleasure Sound Pressures in
Absolute Terms," A. J. Aldridge, 1\ O. E. E. Journal, Oct. 1928, pp. lU-US.
"Effect of the Diffraction Around the Microphone in Sound Measurements," S. Bal-
lantine. Physical Review, Dec. 192S, ])p. 988-992. " Measurements of Sound
Pressure on'an Obstacle," \V. West, hist. Elec. Eng. Journal, 1929, pp. 1137-1142.
CONDENSER AND CARBON MICROPHONES
53
The difference between the pressure and held calibrations is due to
several factors. In the first place the sound is diffracted around the
microphone differently at different frequencies. At frequencies where
the wave-length is large as compared with its external dimensions the
pressure is the same as that of the undisturbed wave. At the higher
frequencies where the microphone is large in comparison with the wave-
length of the sound, the pressure is twice that developed at the lower
frequencies. In the 394-type microphone the effect of diffraction
-46
O db = I VOLT (OPEN CIRCUIT) PER BAR
POLARIZING VOLTAGE = 200 VOLTS
Fig. -i-
lOO 1000
FREQUENCY IN CYCLES PER SECOND
Pressure calibration of the 394 type condenser microphone.
first becomes noticeable in the region of 1200 cycles and reaches a
maximum of 6 db at approximately 2200 cycles. The second factor
which causes a difference between the pressure and field calibrations is
acoustic resonance in the shallow^ cavity in front of the microphone.
This causes the pressure actuating the diaphragm to be higher than
that of the incident sound wave in the frequency region of 1500 to
5500 cycles. The maximum increase in pressure occurs at approx-
imately 3500 cycles. If the sound source is so located relative to the
I -55
100 1000 10000
FREQUENCY IN CYCLES PER SECOND
Fig. 5— Field calibration of the 394-type condenser niicrophone for a direction of
approach of sound normal to the diaphragm.
microphone that the waves approach from a direction normal to the
diaphragm and reflection from surrounding walls and objects is
negligible, the combined effect of diffraction and resonance is to
produce a maximum departure from flatness of approximately 12 db
as is shown by the field calibration Fig. 5.^ If the sound wave travels
9 These curves are taken from unpublished work of P. B. Flanders of the Bell
Telephone Laboratories, Inc.
54
BELL SYSTEM TECHNICAL JOURNAL
along the diaphragm the effective pressure is reduced at the higher
frequencies due to difference in phase. Hence, if the direction of
approach of the sound wave is parallel to the plane of the diaphragm,
the departure from flatness is materially reduced. This is brought
out quite clearly by the field calibration for sound approaching from
a direction parallel to the diaphragm, Fig. 6.'-*
The discrepancy between the pressure and field calibrations of the
condenser microphone involves two important assumptions, namely,
a plane sound wave and no reflection from walls or surrounding objects.
When the microphone is used in a studio much of the sound reaches
the diaphragm by way of reflection from the walls of the room. The
requirement of no reflection is therefore not met and the influence of
the acoustic properties of the reflecting surfaces is added to the char-
acteristics of the microphone. The effect of the diffusion of the
-40
'
~
~
—
n
~
-45
"0 -50
y
\
-
Z
'
— 1
^
10 -55
z
o
Q.
-
\
db= 1 VOLT (OPEN CIRCUIT)PER BAR
POLARIZING VOLTAGE = 200 VOLTS
N
UJ
\
-65
>
\
-
-70
-2P
1000
FREQUENCY IN CYCLES PER SECOND
Fig. 6 — Field calibration of the 394-type condenser microphone for a direction of
approach of sound parallel to the diaphragm.
sound field and the tendency for most materials to be more absorbent
for sounds of high frequency appears to cause the response under
studio conditions to be more nearly like that obtained when the sound
approaches in a direction parallel to the diaphragm and make the
departures from the pressure calibration less marked than the field
calibration for a direction normal to the diaphragm would indicate.
This perhaps accounts in part at least for the instances in which a
corrective network designed to compensate for the field calibration
normal to the diaphragm failed to effect a material improvement in
quality.
The acoustic conditions under which a microphone is used cover a
wide range. It would therefore be difficult if not impossible to adopt
a set of conditions for use in connection with a field calibration of the
condenser microphone, which would be known to be representative
of those encountered in practice. The pressure method of calibration
CONDENSER AND CARBON MICROPHONES 55
on the other hand is definite, simple, and capable of being accurately
duplicated in different laboratories. In view of this situation it would
seem advisable to retain, at least for the present, the thermophone or
pressure method of calibration for general use. In cases where precise
quantitative measurements are required a field calibration of the
microphone should of course be secured under the conditions of actual
use. \^arious methods of making such a calibration have been pro-
posed. The Rayleigh disc has been used extensively in this work
thus far but there are certain very definite limitations to the extent
to which it can be applied. An interesting discussion of the use of
the Rayleigh disc may be found in papers by E. J. Barnes and \V.
\Vest,i« and L. J. Sivian."
It would seem reasonable to expect that future design work would
be directed toward reducing transition, resonance and phase difference
effects to a minimum. The results of work along this line have been
reported by S. Ballantine ^^ and D. A. Oliver.i^ j^ both instances
the mechanical design is such that the resonant cavity in front of the
diaphragm is eliminated and the housing is spherical or streamline
to reduce the diffraction effect. There has as yet been little oppor-
tunity to determine the extent of the practical improvement effected
by these changes in design and the whole discussion continues to be
somewhat academic in character.
Carbon Microphoni<:
Bell's original microphone was essentially a generator and hence
was limited in its output to the maximum speech power available at
its diaphragm. The demand for telephonic communication over
longer distances led to the early introduction of a carbon microphone.
In this instrument the resistance of the carbon element is caused to
vary in response to the sound pressure on the diaphragm and produces
changes in the current supplied from an external source of electrical
potential, which are fairly faithful copies of the pressure changes which
constitute the sound wave. The carbon microphone is therefore in
general an amplifier in which a local source of power is controlled by
the acoustic power of the sound wave.
The carbon element or "button" of the first microphones (Edison,
1877) was made from plumbago compressed into cylindrical form.
^^''The Calibration and Performance of the Rayleigh Disc," E. J. Barnes and
W. West, Inst, of Elec. Eng. Journal, 1927, Vol. 65, pp. 871-880.
""Rayleigh Disc Method for Measuring Sound Intensities," L. J. Sivian,
Philosophical Magazine, March 1928, pp. 615-620.
»2 Contributions from the Radio Frequency Laboratories No. 18, S. Ballantuie,
April 15, 1930. „ ^ ^ ^,.
13 "An Improved Microphone for Sound Pressure Measurements, D. A. Oliver,
Journal oj Scientific Instruments, April, pp. 113-119.
56 BELL SYSTEM TECHNICAL JOURNAL
This t\j)c (){ button was relatively insensitive and shortly after its
introduction the suggestion (Hunnings, 1878) was made that the
space between the diaphragm and the fixed electrode be "partially
filled with puKerized engine coke," '"* in order to increase the number
of contact points and render them more susceptible to the forces
developed by the motion of the diaphragm. When at its best the
Hunnings transmitter was fairly efiicient but at times was erratic in
its performance due in part to the nature of the microj^honic material.
In 1886 P2dison ^^ proposed the use of granules of hard coal Avhich had
been heat treated. This was an important advance, for carbon made
from anthracite coal is used not only in the microphones which are
being considered in this paper but in commercial telephone trans-
mitters as well.
As in the case of the condenser microphone, the displacement of the
diaphragm of the carbon microphone must be substantially constant
at all frequencies if uniform response is to be obtained. In the early
microphones of the carbon type, diaphragm resonance introduced
rather prominent irregularities in response. Air damped stretched
diaphragms offered one solution of this problem. During the World
War instruments of this type were developed and applied to the
problem of locating airplanes. In 1921 double button stretched
diaphragm microphones were made available for use with the public
address equipment installed for the inaugural address of President
Harding and the excercises at Arlington on Armistice Day.^^ The
carbon microphones employed in sound picture recording are of the
stretched diaphragm double button type. The electrical output
from this type of microphone is not only of substantially uniform
intensity over a wide frequency range but due to the "push-pull"
arrangement of the buttons is comparatively free from harmonics.
A typical example of the present day carbon microphone is shown in
the photograph, Fig. 7. F'ig. 8 is a cross-sectional view of the same
type of microphone.
The diaphragm is made from duralumin. .0017 inch in thickness and
is clamped securely at its outer edge. The clamping surfaces are
corrugated and emery cloth gaskets are provided to prevent slipping.
The stretching of the diaphragm is done in two steps. The initial
stretching ring is first advanced by means of six equally spaced screws
until the diaphragm is smooth and free from irregularities. The inner
or final stretching ring is then adjusted to a position which gives the
1^ "Beginnings of Telephony," F. L. Rhodes, p. 79, 1929.
«U. S. Patent Xo. 406,567, 1889.
"> "Public .Address Svstems," 1. W. (".rtn-n and |. I'. .Maxtield, .1. /. E. E. Journal,
.\pril 192.^ pp. .U7-358.
CONDENSER AND CARBON MICROPHONES
57
diaphragm a resonant frequency of 5700 cycles per second. The
method employed in making the determination of the resonant
frequency is substantially the same as that used in connection with the
assembly of the condenser microphone, with the exception that the
Pier. 7_Westeni Electric Company's 387-type carbon microphone.
DIAPHRAGM
FINAL
STRETCHING
RING
DAMPING
PLATE GROOVE
INITIAL
STRETCHING -•
RING
Fig. 8— Cross-sectional view of the 387-type carbon microphone.
frequency at which the ma.ximum output occurs is usually determined
by ear rather than by the coupler method previously described.
In order to insure a uniformly low contact resistance the portions of
the diaphragm which are in contact with the granular carbon are
covered with a film of gold deposited by cathode sputtering.
58
BELL SYSTEM TECHNICAL JOURNAL
A spacing washer .001 inch in thickness separates the diaphragm
from the damping plate. A single concentric groove is provided in
the damping plate.
The buttons are of the con\entional cylindrical type but are provided
with a novel form of closure to jirevent carbon leakage at the point
where they make contact with the diaphragm. The closure consists
of twenty-seven rings of .0004 inch paper clamped firmly together at
the outer edge and spreading apart at the inner edge to form a structure
which effectively seals the junction between the diaphragm and the
buttons without adding materially to the mechanical impedance.
As has already been pointed out the granular carbon is made from
selected anthracite coal. The size of the granules is such that they
will pass through a screen having 60 meshes per inch but will be re-
tained on a screen having 80 meshes per inch. Before heat treatment
the raw material is treated with hydrofluoric and hydrochloric acids
to reduce the ash content. Each button contains .060 cc. of carbon,
i.e., about 3000 granules.
The bridge which supports the button on the front of the diaphragm
partially closes the acoustic cavity on that side. It is essential,
therefore, that it be so proportioned as to have a minimum reaction
on the response of the microphone and yet provide the required degree
of rigidity. It was this consideration that led to the smooth stream
line contour now employed.
-40
-45
(0-50
(0 -55
g
^
"^
N
—
-
^
J
s
â– V-
f
4
db=l VOLT (OPEN CIRCUIT) PER BAR
2
_
1
50
10.000
FREQUENCY IN CYCLES PER SECOND
Fig. 9 — -Pressure calibration of the 387-type carbon microphone.
Referring to Fig. 9 it will be observed that the adoption of an air
damped stretched duralumin diaphragm for the carbon microphone
has resulted in an instrument having a substantially uniform response
over a wide range of frequencies. The arrangement of the apparatus
employed in securing the data from which this curve was plotted is
shown in the photograph, Fig. 10. The microphone under test was
mounted in a highly damped room at a distance of six to eight feet
from a source of sound which consisted of two loud speaking receivers.
CONDENSER AND CARBON MICROPHONES
59
One of the receivers was the conventional form of moving- coil direct
radiator and was used to provide sound in the lower frequency range.
The other was a special moving coil receiver with a short horn so
designed as to serve as an efficient source of sound up to 10,000
cycles.i^ To reduce the effect of standing waves the mounting for the
receivers was so constructed that they could be rotated through a
circle approximately five feet in diameter and always face the micro-
phone under test. Before starting the test of the carbon microphone
the receivers were calibrated by placing a calibrated condenser micro-
pig_ 10 — Apparatus employed in calibrating the 387-type carbon microphone.
phone at the point where the test instrument was to be located and
determining the receiver current required to produce a pressure of one
bar (one dyne per square centimeter) on the microphone diaphragm.
The condenser microphone was then removed and the test microphone
substituted. The open circuit voltage developed by the microphone
when supplied with a direct current of .025 ampere per button was then
measured. The data obtained in this way are essentially a "pressure
calibration" of the microphone and in interpreting them in terms of
"field" performance the same factors must be taken into account
17 "An Efficient Loud Speaker at the Higher Audible Frequencies," L. G.Bost-
wick, Journal of the Acoustical Society, Oct. 1930, pp. 242-250.
60
BELL SYSTEM TECHNICAL JOURNAL
which have been discussed in considerable detail in connection with
the condenser microphone.
The circuit employed in measuring the response of the carbon micro-
phone is shown on Fig. 11. Two steps are involved in the calibration
of the sound source. With the output terminals of the microphone
circuit and the sound source short circuited and the polarizing voltage
for the condenser microphone removed, the attenuator is adjusted
until the voltage applied to the measuring circuit is that developed by
the condenser microphone when a sound pressure of one bar is im-
pressed on its diaphragm. A record is made of the reading of the
cni .MH CONDENSER
c
HIGH PASS
FILTER
60 CYCLES ANDn
135 CYCLES
POTENTIAL
ATTENUATOR
CARBON
MICROPHONE
LOW
PASS
FILTER
X
_
-o
THERMO-
COUPLE
^"X
THERMO-
-COUPLE
'C^
RECEIVER AND
ATTENUATOR
CURRENT
METER
>;
POWER
OSCILLATOR
Fig. 11 — Circuit emi)loyecl in calil)raling the 387-type carbon microphone.
output meter in the measuring circuit. The polarizing voltage is
then applied to the condenser microphone. After the output terminals
of the attenuator have been short circuited an alternating current of
a known frequency is supplied to the sound source and the magnitude
of this current adjusted until the meter reading is the same as that
previously obtained with the attenuator. This completes the cali-
bration of the sound source for that frequency. After the carbon
microphone has been placed in the position previoush- occupied by
the condenser microphone, the polarizing \oltage is once more removed
from the condenser microphone and the output from the carbon
microphone circuit impressed on the measuring circuit. The reading
CONDENSER AND CARBON MICROPHONES 61
of the output meter is recorded. The sound source and carbon
microphone circuit are then short circuited and the output from the
attenuator again applied to the measuring circuit. The attenuator
is adjusted until the reading of the output meter is the same as was
previously obtained with the carbon microphone in circuit. In this
way the voltage applied to the measuring circuit when the carbon
microphone is in operation is determined. The open circuit voltage
developed by the carbon microphone may then be computed from the
voltage and the constants of the microphone circuit. At the locations
where these measurements were made a certain amount of interference
from 60-cycle circuits and low frequency acoustic disturbances was
encountered. The high-pass filter in the measuring circuit was intro-
duced to facilitate the measurements under these conditions. The
adjustable low-pass filter was used to confine the measurements to
the fundamental frequency. Only that portion of the apparatus to
the left of the dotted line was mounted in the damped room.
The two buttons of the carbon microphone are identical in their
dimensions and if the granular carbon is in the same mechanical state
have substantially the same electrical characteristics. They are also
practically free from the cyclic variations in resistance known as
"breathing" which result from the temperature changes caused by
the power dissipated in the granular carbon. It is, however, a matter
of every day experience that a given mass of granular material will
occupy different volumes, depending upon the configuration of the
particles. In the case of microphone carbon this change in configura-
tion of the granules results in changes in the contact forces of sufficient
magnitude to affect the resistance and sensitivity. If these changes
occur in unequal amounts in the buttons electrical unbalance results.
When complete balance exists the electrical output is free from all
harmonics introduced by the circuit. Hence, in using the microphone
care should be taken to see that a fair degree of balance between the
buttons is maintained.
The performance of a carbon microphone may be affected adversely
by cohering of the granules. Severe cohering is accompanied by a
serious reduction in resistance and sensitivity which persists for an
extended period unless the instrument is tapped or agitated mechan-
ically. One of the common causes of cohering is breaking the circuit
when current is flowing through the microphone. Experiment has
shown that the insertion of a simple filter consisting of two .02 mf.
condensers and three coupled retardation coils each having a self-
inductance .0014 henry, will effectively protect the microphone button
from cohering influences without introducing an appreciable trans-
62 BELL SYSTEM TECHNICAL JOURNAL
mission loss. This filter may be located in the base of the mounting
or in a container fastened to the back of the microphone.
Aging of granular carbon may result from changes in the contact
surface caused either by mechanical abrasion or overheating due to
excessive contact potentials. Aging is usually accompanied by an
increase in resistance and loss in sensitivity. Care should therefore
be exercised in the use of the carbon microphone that it is not sub-
jected to unnecessary vibration which would cause the granules to
move relative to one another and abrade the surfaces. The use of
abnormally high voltages should also be avoided.
The quality of transmission obtained with the double Initton carbon
microphone compares favorably with that secured with a condenser
microphone. The carbon microphone also requires less amplification.
There is, however, one characteristic which limits its use, namely
carbon noise. The level of the noise is much higher than that due to
thermal agitation within the carbon granules ^^ and appears to be
caused by heating at the contacts between the granules. A certain
amount of gas is contained in the pores in the contact surfaces. When
current passes through the button, a sufiicient increase in contact
temperature takes place to cause a portion of this gas to be driven off
and produce the non-periodic changes in resistance which give rise
to carbon noise.
In conclusion it may be stated that the condenser and carbon types
of microphones have been developed to a point where there is little to
choose between them from the standpoint of quality of transmission.
The design from a mechanical standpoint has also been carried to a
point where little difficulty should be experienced in their use if reason-
able precautions are exercised. Although requiring less amplification
than the condenser microphone the extent to which the carbon micro-
phone is used at present is limited by the higher noise level obtained.
The condenser type of microphone has therefore been adopted for
most of the recording work in the sound picture field.
'* "Thermal Agitation of Electricity in Conductors," J. B. Johnson, Physical
Review, July 1928, pp. 97-109.
Certain Factors Affecting the Gain of Directive Antennas*
By G. C. SOUTHWORTH
This paper analyzes the performance of antenna arrays as influenced by
certain variables within the control of the designing engineer. It starts with
an extremely simple analysis of the interfering effects produced by two
sources of waves of the same amplitude. This is followed by a short dis-
cussion of a paper by Ronald Foster, which considers two antennas and also
16 antennas when arranged in linear array. Two antennas separated in
space by J^ wave-length and in phase by i^ period give sensibly more
radiation in one direction than in the opposite. This, for convenience, has
been called a unidirectional couplet. A number of these couplets may be
arranged in linear array, thereby giving an extremely useful directive
system. Diagrams are shown for such arrays as affected by the number and
spacings of the indi\'idual couplets. The gains from such arrays are
calculated and data are given showing fair agreement between calculation
and observation.
Directional diagrams for arrays of coaxial antennas indicate that some-
wliat less gain may be expected from this form than when the elements are
spaced laterally. Combinations of tliese two types of arrays give marked
directional ]3ro])ertios in both their horizontal and vertical planes of refer-
ence. This principle lias been used ratiier generally in short-wave coni-
nmnication. This paper also discusses effects resulting from combining
two or more arrays. In one case the sjiace between two arrays tends to
emphasize sjiurious lobes. The directional diagram of such a combination
may be rotated within limits by changing tlie phasing between adjacent
arrays or sections of an array. In all of the above cases the influence of
the earth is ignored.
A mathematical appendix gives general equations for calculating di-
rectional diagrams of linear arrays. Special cases of these equations apply
to the figures included in the main part of the text. General equations are
also given for calculating the gains of arrays. Similar equations permit the
areas of diagrams to be calculated. An extended bibliography on antenna
arrays is appended.
Introduction
THROUGHOUT the development of radio communication the
engineer has aspired to a directive system whereby radiation
might be projected from one point to another with a maximum
of efficiency and a minimum of interference with adjacent stations.
Also, he has aimed at similar directivity at the receiver to improve
the signal-to-noise ratio and otherwise discriminate against un-
desirable signals. It was recognized at a very early date that directive
radio based on wave interference was feasible provided sufficiently
short waves could be utilized, and as a result many interesting sug-
gestions to this end were made. However, as is well known, the early
development of the radio spectrum proceeded in the direction of long
* Presented at Convention of I. R. E., Toronto, Ont., Canada, Aug. 19, 1930.
Proc, I. R. E., Sept. 1930.
63
64 BELL SYSTEM TECHNICAL JOURNAL
waves rather than short waves, thereby deferring many of the appHca-
tions of these suggestions.
The principle of wave interference on which most short-wave
systems of directive radio are based has probably been known for
several centuries. However, the first thorough treatment of this
subject was by Sir Thomas Young, ^ who, together with Fresnel,
securely established the wave theory of light in the early part of the
last century. Even Hooke and Huygens, who had offered the wave
theory over a century earlier, failed to recognize the full significance
of interference.
When Hertz started his celebrated experiments to verify Maxwell's
theory he was, of course, in full knowledge of these phenomena and
their explanation, and invoked their use in proving the existence of
electric waves. It is interesting that in some of his experiments he
made use of parabolic mirrors for both transmitting and receiving,
having directional characteristics very similar to those sometimes
used in present day radio practice. It is also of interest that he found
that parallel wires stretched over a frame were quite as eff^ective as a
reflector as a continuous sheet of metal of similar dimensions, pro-
vided the wires were kept parallel to the lines of electric force of the
arriving wave. He apparently did not in\estigate the effect of varying
the spacing nor the length of the parallel wires, nor did his subsequent
experiments otherwise tend toward the present day antenna array
technique.
This paper treats in an elementary way certain aspects of the
antenna array problem, principally as regards the manner in which
calculated directivity is affected by the number and spacing of the
individual antennas which go to make up the array. The theory is
applicable only to those forms of directive antennas which may be
resolved into a series of individual sources. It does not apply to the
so-called wave antenna. However, principles are included which have
for some time been in general use in combining two or more such
antennas.
Extensive study has been given to directive antenna systems for
use in transoceanic radiotelephony. Papers dealing with this general
subject have appeared from time to time.' F'urther work is in prog-
ress. Papers by E. J. Sterba and also by E. E. Bruce and H. T. Friis of
the Bell Telephone Laboratories are in preparation which will include
1 Phil. Trans, of Royal Soc, 92, 12; 1802.
^ R. M. Foster, "Directive diagrams of antenna arrays," Bell Sys. Tech. Jour.,
292, May, 1926. Austin Bailey, S. W. Dean, and \V. T. Wintringham, "Receiving
system for long- wave transatlantic radiotelephony, Proc. I. R. E., 16, 1694, December,
1928. J. C. Schelleng, "Some problems in short-wave radiotelephone transmission,"
Proc. I. R. E., 18, 913; June, 1930.
GAIN OF DIRECTIVE ANTENNAS
65
certain calculated data similar to those contained in the present paper,
and also experimental results obtained from tests on actual antennas of
various sizes and proportions.
In the early part of the following discussion each antenna is con-
sidered as a spherical source of waves which radiates equal power in
all directions. Furthermore, it assumes that the current in each
(a)
Fig. 1— Interference pattern. Two equiphased sources spaced one-half wave-length.
individual source, in a given array, is the same and is not materially
affected in either magnitude or phase by its proximity to other sources.
The fair approximation to which these calculated results are realized
in practice bespeaks the justification of these assumptions.
The various steps by which present day directional radio has been
developed are extremely interesting, but they are so involved in the
development of radio itself that their enumeration is considered out-
66 BELL SYSTEM TECHNICAL JOURNAL
side the scope of this paper. However, bibliographies are cited below
covering some of their important phases.
Elementary Principles
The interference patterns resulting from a number of individual
sources of waves, such as antennas, are dependent on both their
spacial arrangement and the magnitudes and relative phases of their
forces. This makes possible an almost unlimited number of com-
binations of which only a portion have thus far found use in com-
(a)
Fig. 2 — Interference pattern. Two sources separated in space by one-fourth wave-
length and in time by one-fourth period.
munication. This paper will restrict itself mainly to some cases which
are already finding general application. As a suitable introduction
to this subject, a very simple case of wave interference is discussed in
the following paragraph.
Figs, la and 2a depict in a rough way the interference resulting
from two independent sources of spherical waves of the same ampli-
tude. In the first case they are spaced ^ wave-length but are assumed
to be oscillating in phase. In the second case the two sources are sepa-
rated in space by 3^ wave-length and in phase by ]i period. Crests
GAIN OF DIRECTIVE ANTENNAS 67
and troughs are represented respectively by solid and dotted lines.
At points where either two crests or two troughs arrive simultaneously
the resultant wave is greatly enhanced, whereas at certain other points
crests and troughs arrive together, thereby neutralizing each other's
effects. At certain intermediate points these interfering effects are
only partially complete. Accompanying each figure is a directive
diagram (lb and 2b), plotted in polar coordinates, which shows the
effectiveness of the wave in each direction. The circle drawn outside
each diagram indicates the effect if the radiation had proceeded from
a single non-directional source similar to each of the above. The
ratio between the areas of the circle and the inscribed diagram gives
roughly the power improvement of such a device as manifested in the
intensity of the radiated wave. A more exact calculation of this
improvement requires an integration of the force components over a
unit sphere.
Linear Antenna Arrays
Most directive antenna systems now in general use for short
waves may be regarded as special applications of the linear array.
This type consists of two or more antennas all having currents of equal
amplitude, equispaced along the same straight line. The properties
of such arrays have been treated very generally by Foster,^ whose
paper included several hundred directive diagrams, taken in a bi-
secting plane perpendicular to the axis of each antenna of the array,
and typical of the results which may be expected from two antennas
and from arrays consisting of 16 antennas. A portion of these dia-
grams have been reproduced in Figs. 3 and 4 below. The same
principles are applicable to both transmission and reception.
In Fig. 3 are shown diagrams resulting from two antennas as the
separation is increased from to 1 wave-length in steps of 3^ wave-
length and the phase increased from to >2 period in steps of }4
period. The line or axis of the array is assumed to be horizontal and
the specified phase difference is such that the current in the right-
hand antenna is lagging for a transmitting system and leading for a
receiver. It will be noted that for phase differences of both and }4T
the diagrams are symmetrical about both the horizontal and vertical
axes of the figure, whereas for other phases the figures are asymmetrical
about the vertical axis except for certain limiting cases. Of these
asymmetrical diagrams, that corresponding to phase and spacial sepa-
rations both of I'i (Fig. 3b) is of particular importance and forms
the basis of the so-called reflector effect. This particular combination
of two sources is referred to later as a unidirectional couplet.^ In
2 Loc. cit.
* In this, and in other cases in this paper, radiation is referred to as unidirectional
when sensibly more power is propagated in one direction than in others.
68
BELL SYSTEM TECHNICAL JOURNAL
"^
Q. a>
rtf-c
^ -T-1
C.2
2 a
in a>
c
â– 5
CAIN OF DIRECTIVE ANTENNAS 69
passing it is also of interest to note that the diagram of the coil or
frame aerial as generally used is intermediate between Figs. 3c and
3d. Its diagram would not differ essentially from its neighbors, Figs.
3d, 3e, or 3f , except for scale. This scale may conveniently be regarded
as a measure of the impedance of the device, or possibly its radiation
efficiency, but not necessarily a measure of its usefulness.
Fig. 4 shows similar diagrams resulting from 16 antennas for vari-
ous phase and space relations. As in Fig. 3, diagrams in the top and
bottom rows corresponding respectively to phases of and K^ T are
symmetrical about both the horizontal and vertical axes. The dia-
grams in the top row are in general bidirectional, while the bottom row
has one bidirectional diagram corresponding to phase and space
differences both equal to >^. It is of interest that for the most part
cases where the phase and space separations are numerically equal
correspond to unidirectional diagrams. However, these diagrams are
only moderately sharp and thus far such arrays have not been used
extensively in practice.
Referring again to the diagrams in the top row corresponding to 16
antennas all driven in phase, we note that directivity becomes progres-
sively sharper as the spacing is increased until in the vicinity of 15/16X
appendages develop which soon surpass in magnitude the desired lobes.
This effect is present in the commercial array, and limits, as we shall
later see, the gain that may be derived from a given number of elements.
The diagrams shown in Fig. 4 for 16 antennas are typical of others
where the number of antennas in linear array is fairly large.
The Linear Array and Reflector
One type of array now in commercial use consists of two parallel
linear arrays of equiphased elements where the two parallel arrays are
spaced M wave-length and differ in relative phase by }/ii period. It is
convenient to regard such a device either as two independent linear
arrays, each having a directional characteristic as shown in the top
row of Fig. 4, or as an array of couplets, each couplet of which has by
itself a heart-shaped characteristic. Both antennas of the couplet may
be independently driven at their prescribed phase separation of 3i
period, or one may derive its power from that radiated by the other, in
which case the proper phase relation is automatically approximated •*
and the same practical result is obtained. In the latter case one is
^ The problem of the reflecting antenna has been considered by Wilmotte and
McPetrie, Jour. I. E. £., 66, 949, Englund and Crawford, Proc. I. R. E., 17, 1277;
August, 1928, and Palmer and Honeyball, Jour. I. E. E., 67, 1045. Their conclusions
indicate that the optimum separation between a single antenna and its reflector to
give maximum forward radiation is roughly X/3. However, it appears that when
several antennas and reflectors are involved a separation more nearly X/4 is optimum.
70 BELL SYSTEM TECHNICAL JOURNAL
frequently known as the driven antenna and the other the reflector.
This viewpoint is perhaps only a convenience and may not be al-
together correct. An array of the above type transmits and receives
best in a direction at right angles to its principal dimension. This
type is, therefore, frequently known as a broadside array.
Directive Diagrams from Arrays and Reflectors
In Fig. 5 is plotted a series of diagrams in a bisecting plane normal
to the axis of each antenna of the array for different broadside arrange-
ments such as are used commercially. They are systematically ar-
ranged horizontally in the order of the number of couplets in the array,
and vertically with the increased spacing between adjacent couplets.
Several different forms of such directive diagrams are possible,
which may be plotted in either polar or rectangular coordinates. In
one form all diagrams are roughly of constant area and relative gains
from various antenna systems are expressed in terms of the principal
radius vector. In the second form the length of the principal radius
vector remains constant and the relative gain is roughly inversely
proportional to the area of the diagram. The second of these forms has
been adopted in this paper largely because of the relative simplicity of
the equation of the diagram and the facility with which properties of
antennas may be determined.
In the lower left-hand corner of Fig. 5 will be found a plan showing
the arrangement of the elements relative to the important direction of
transmission. At its right is the general equation of these diagrams.
This formula is also given as equation (14) of the appendix where the
analytical theory of arrays is developed. Below each diagram is the
ratio of the area of the circumscribed unit circle to the area of the hori-
zontal diagram. Here also will be found the ratio of the area of the
subordinate loops to the area of the main loop. The total area may be
measured approximately with a planimeter or calculated more accu-
rately by equation (32) in the mathematical appendix. In making up
Fig. 5 each diagram was accurately plotted on standard polar coordin-
ate paper from perhaps a hundred calculated points. This was then
reduced photographically and the several diagrams were assembled.^
Inspection of the diagrams shows that increasing the number of
couplets increases in all cases the sharpness of the main loop and
hence the gain of the array. However, increasing the separation be-
* The diagrams used in this paper were calculated by a group of the Department
of Development and Research of the American Telephone and Telegraph Company,
under the direction of Miss E. M. Baldwin. Most of the material was checked by
Mrs. Isabel Bemis, who assembled it in its present form and prepared the attached
bibliography.
lA SA
TS
V\(V)
^ i'X sh i^ 4?* |A aX jA fi\ lA
lA *A SA iA §A ^A ^ |;^ g?, ^a lA |A iA fA SA |A HA {fA lA A
O(S)(D(D(D(D(D(D00(D000(D000(Da)0®00(D00(D(D®®
- o
©000000000000000000000000000000
U ( O )( 8 )(Q
00©
O0©©
0©©©©
:;©©©©
. no '^^
•^000
I^ig- 4 — Directive amplitude diagrams for an array of si
e-Iengths (X) along the top. Phase difference in periods (T) at the left.
r\
S=74.0
S = I09.2
5 = 130.3
— ARRANGEMENT OF ARRAY—
-U .
AX
t DIRECTION
• •- *- OF
TRANSMISSION
— NOT ES —
AREA OF UNIT CIRCLE TO THAT OF DIRECTIONAL DIAGRAM
AREA OF SUBORDINATE LOOPS TO THAT OF MAIN LOOP
OF WAVE LENGTH SPACING BETWEEN ELEMENTS
COLUMN
— ARRANGEMENT OF ARRAY—
AX
• • —
NOT ES —
S=74.0
S = I09.2
5 = 130.3
AREA OF UNIT CIRCLE TO THAT OF DIRECTIONAL DIAGRAM
AREA OF SUBORDINATE LOOPS TO THAT OF MAIN LOOP
OF WAVE LENGTH SPACING BETWEEN ELEMENTS
DIRECTION tOLUMN
â– *â– OF
TRANSMISSION
— ARRANGEMENT OF ARRAY —
— EQUATION OF DIAGRAM
SIN (nttA sin 0)
N SIN(irA SIN t]
COS ^ (cos 0-1)
S = RATIO OF AREA OF UNIT CIRCLE TO THAT OF DIRECTIONAL DIAGRAM
R = RATIO OF AREA OF SUBORDINATE LOOPS TO THAT OF MAIN LOOP
TRANSMISSION
Fig. S — Horizontal plane diagrams — number of couplets
separation in wave-lengths.
GAIN OF DIRECTIVE ANTENNAS
71
tween couplets increases the gain only up to a certain point, after
which the formation of parasitic lobes decreases the effectiveness of the
array. The trend of these gains may be illustrated more effectively in
graphical form.
In Fig. 6 calculated gain ratio is plotted against number of couplets
giving one graph for each separation considered. These ratios are not
based on the data given in Fig. 5, but were obtained from the integra-
tion of the equation of the directional diagram over an arbitrary
sphere by use of equation (27) below. It may be noted that for many
conditions the difference between these methods of calculating gain is
only moderate. These power ratios are for the most part linear,
130
120
110
100
90
O 80
cc
70
iti 60
§
O 50
a.
40
30
20
10
y
y
y
8/
y
/
X
y
/
A
/•
y
y
/'
/
4-K
4
y
y
y
/
^'
y
^
/ ,
A
^
-^^-^
-"
"
/
X
y^
^ --
^ "^
,^
/
/
X
\^
^-
—
^
---
—
-z'
/'^
/
y
^-'
-^
"
^
^
X
-^
-^
4
^
^
X
^^^
^
^
^
12 16 20 24 28 32 36 40
NUMBER OF COUPLETS
Fig. 6 — Antenna arrays. Calculated power ratios vs. number of couplets.
indicating that such gains are proportional to the length of the array.
This is in keeping with the view that a receiving antenna can intercept
wave power more or less in proportion to its dimensions. It is also
interesting to note that the slope of the curve of X/2 is approximately
twice that for X/4, so that 16 couplets spaced }i wave-length give
approximately the same gain as eight couplets spaced Yi wave-length.
This again shows that the length of the array is the most important
criterion in determining its gain. In Fig. 7 the same data have been
plotted in decibels.
In Fig. 8 gains expressed in decibels are plotted against the separa-
tion between elements. This shows more definitely the trend of the
CAIN OF DIRECTIVE ANTENNAS
71
tween couplets increases the gain only up to a certain point, after
which the formation of parasitic lobes decreases the effectiveness of the
array. The trend of these gains may be illustrated more effectively in
graphical form.
In Fig. 6 calculated gain ratio is plotted against number of couplets
giving one graph for each separation considered. These ratios are not
based on the data given in Fig. 5, but were obtained from the integra-
tion of the equation of the directional diagram over an arbitrary
sphere by use of equation (27) below. It may be noted that for many
conditions the difference between these methods of calculating gain is
only moderate. These power ratios are for the most part linear,
130
y
y
y
7>
8/
no
/
y
/
•
/â–
.^
^
/'
/
4
>
y
O 80
1-
/
• '
,^
^
/ ,
Y
^
^ -y-
^--
''
u 60
O 50
/
^
^
^ ^
-"
,-
/
/
^
X
\^
^^
'^
^
--
"
40
^y
7"—
/
X
^
^-'
^â– ^
â– ^
30
/^
^
* —
^
^
â– ^
-^
4
20
^
^
^
-^
10
%
3
2
NUN/
6
BER
2
OF
CO
2
UPL
4
ETS
'
8
â– ;
2
-
6
4
Fig. 6 — Antenna arrays. Calculated power ratios vs. number of couplets.
indicating that such gains are proportional to the length of the array.
This is in keeping with the view that a receiving antenna can intercept
wave power more or less in proportion to its dimensions. It is also
interesting to note that the slope of the curve of X/2 is approximately
twice that for X/4, so that 16 couplets spaced \i wave-length give
approximately the same gain as eight couplets spaced yi wave-length.
This again shows that the length of the array is the most important
criterion in determining its gain. In Fig. 7 the same data have been
plotted in decibels.
In Fig. 8 gains expressed in decibels are plotted against the separa-
tion between elements. This shows more definitely the trend of the
72
BELL SYSTEM TECHNICAL JOURNAL
antenna gain to a maximum, after which spurious lobes become of
importance. Fig. 8 suggests that the spacing, giving optimum gain,
would be the desideratum in antenna design. However, this is not
24
22
20
18
UJ
? 14
U
LU
Q 12
Z
- 10
z
< 8
o
6
4
2
_-
TK
8^
_^ --
--
__-^-
-• ^^
--
- —
— ■"
>
2 â–
\
3>
4
V
^
â– ^
"^
^ ^
'-
— -
-" *"
_ _
.)>
r"
J^
^-
â– -^
^-
--
— "
'"
_-
--
,^^
^
i^
^^
-^
1^
^
— "
•^
/
^
^
^
^
^
4
/
'a
^
^
//.
K/
/
V
/
/
16 20 24
NUMBER OF COUPLETS
Fig. 7 — Antenna arrays. Calculated gains vs. number of couplets.
22
20
0.^25^
"
^
^\
« 1 6
lU 14
^^
^^
Ji.iiS-
\
^^^
^^
-^
_^
^^^
/^
y^
^^^
,— — *' '^
N\
::ii2
Q
'/-
^
---^
^^
â–
^
^ \
? 8
<
/
^;^^
^^
^^
^^
^^
//
y^ ^
^^'
^^
"
~~-
I /y
/y^
^^^
-^
'^==^-^
0.3 0.4 0.5 0.6 0.7
FRACTIONAL WAVELENGTH SPACING
Fig. 8 — Antenna arrays. Calculated gains vs. lateral spacing between couplets.
necessarily the case, as we shall presently see. It has already been
pointed out that the over-all length of array, rather than the spacing
or the number of conductors per unit length, constitutes the most
GAIN OF DIRECTIVE ANTENNAS
73
important factor in determining the gain. Furthermore, minimum
area diagrams are frequently attended by fairly large spurious lobes
which are undesirable particularly on receiving antennas. Also the
no
^
^y'
100
^
^^
90
9 80
<
a 70
<
t
^
^
^
>^
o
^
X
50
z
y
^
<
> 30
^
y^
^
^
y^
UJ 20
^
^^
10
6 8 10 12 14
LENGTH OF ARRAY IN WAVELENGTHS
Fig. 9 — Approximate gains to be expected from arrays of couplets for spacings of
approximately X/4 and X/2.
'^Or
" "
.--
.'-'
^^
^'-
,"'
--
--''
,^
^
^
-J
y
^
IB 15
/
y
a 14
z
II
/
/
/
/
/
/
10
9
8
/
/
2 4 6 8 10 12 14 16 18 20
LENGTH OF ARRAY IN WAVELENGTHS
Fig. 10 — Approximate gains to be expected from arrays of couplets for spacings of
approximately X/4: and X/2.
cost of an antenna system of a given height is more or less proportional
to its length, and in many cases is not materially affected by the number
of conductors present. These considerations, together with the fact
74
BELL SYSTEM TECHNICAL JOURNAL
that proper phases may often be most readily accomplished with
intervals of either 34 wave-length or }4 wave-length, have led to a
rather general adoption of these closer spacings.
In Fig. 9, approximate gain ratios from arrays of various lengths
have been plotted. These are most applicable for separations in the
vicinity of ]i and l4 wave-length. Fig. 10 shows the same data
plotted in decibels. Within these limits, it appears that the gain ratio
may be expressed by the simple formula G = KL, where L is the array
length in wave-lengths and K is approximately 5.6. The result
expressed in decibels is G' = 10 \ogiQ{KL).
Measured Antenna Gains
The degree to which the gains calculated above are approximated in
practice is indicated by the data given in the diagrams of Figs. 11 and
12 and in Table I.
TABLE I
Array
Desig-
nation
Nominal
Operating
Frequency
Megacycles
Number
Couplets
Spacing
Measured
Gain Over
Similar
Single
Element
db
Calculated
Gain
db
Differ-
ence
db
1-A
2-A
3-A
1-B
2-B
3-B
4-B
2-C
3-C
1-C
D *
18
18
18
12
12
15
15
10
10
9
14
24
24
24
24
24
24
24
24
24
18
9
X/4
X/4
X/4
X/4
X/4
X/4
X/4
X/4
X/4
X/4
X/2
15.3
15.2
15.0
15.6
14.5
13.6
16.6
16.3
15.5
13.6
13.0
15.0
15.0
15.0
15.0
15.0
15.0
15.0
15.0
15.0
13.8
13.7
-f 0.3
-i-0.2
0.0
+ 0.6
- 0.5
- 1.4
+ 1.6
+ 1.3
+ 0.5
- 0.2
- 0.7
* This antenna actually consisted of two arrays of four couplets each spaced
laterally by one wave-length. The resultant diagram of such an array is for all
practical purposes the same as that produced by a continuous array of nine couplets.
Fig, 11 shows a calculated diagram corresponding to certain
receiving arrays used in the transatlantic telephone service between
America and England. Several points are plotted on this diagram
which correspond to the relative strengths of signals received at vari-
ous angles. These points were obtained by observing the relative
received signal voltage, measured on a standard field-strength measur-
ing set connected to the array as an electric oscillator of constant
amplitude was carried around the array at a distance of perhaps 20
wave-lengths. The plotted data correspond to the case where the
GAIN OF DIRECTIVE ANTENNAS
75
reflector was "floating." Although this arrangement most nearly
corresponds to the conditions assumed in the calculated curve, it is not
necessarily the most desirable adjustment to minimize noise arriving
from the rear. This diagram corresponds to the antennas designated
as 1-A, 2-A, and 3-A in Table I. These antennas consist effectively of
24 vertical couplets spaced horizontally at intervals of 14 wave-length.
In this table are given further data on the strength of signals
received on arrays, as compared with those received simultaneously on
a single element of similar structure and height above earth. The
different antennas represented involve varying conditions of wave-
Fig. 11 — Calculated directional diagram. Twenty-four couplets spaced one-fourth
wave-length. Circles indicate experimental points.
length, height above earth, adjacent terrain, and types of support.
These details are not believed to be of sufficient importance for dis-
cussion here. Two different array lengths are represented. The rela-
tive gains were substantially the same when observed on a local source
of waves and when the signal came from a distant station. The last
array represented in Table I was one used for transmitting. To effect
the test, equal power was transmitted alternately from the array and
from a single element while comparative measurements of electric
field strength were made at a distance of approximately 3500 miles.
The datum given is the mean of perhaps 100 observations extending
over a total of eight hours on three different days. Two errors are
76
BELL SYSTEM TECHNICAL JOURNAL
involved in the data of Table I. One is due to the doubtful magnitude
of a correction necessary to account for the various heights at which
the arrays were located above the earth and the second is the error of
measurement of gain as compared with the reference antenna. These
errors are approximately equal and together amount to ± 1 db.
In order to test further the agreement between measured gains and
those calculated from the simple assumptions above, a receiving array
was assembled step by step and corresponding measurements made.
Certain precautions, such as to maintain impedance matches at points
of coupling, were observed. The resulting data were plotted as points
in Fig. 12. A smooth curve represents the corresponding calculated
16
^
12
-1 10
UJ
m
o
'
^^
â– ^
D
z
? 6
<
O
4
2
n
/
/
•
2 4 6 8 10 12 14 16 le 20 22 24 26
NUMBER OF COUPLETS
Fig 12 — Relation of measured to calculated gain of receiving antenna array at
14,350 kc.
data. It will be observed that the measured values are consistently
higher than those calculated at the lower end of the curve, and in this
region the agreement can hardly be regarded as satisfactory. How-
ever, limited time prevented a thorough study of the errors of measure-
ment. Consequently these limited data may not be regarded as any
adequate test of the theory.
Combinations of Arrays
It may be shown that two or more similar directive systems may
be combined to give a total directive effect, represented by the product
of the individual effect, multiplied by the group effect. This principle
is partially covered by equation (35) of the mathematical appendix.
GAIN OF DIRECTIVE ANTENNAS 77
Two cases are of special interest. First, it is sometimes desirable to
divide an array into two or more bays, in order to make room for a
supporting structure. This, of course, gives rise to a definite discon-
tinuity in the over-all array.
Fig. 13 shows a series of diagrams resulting from a typical case
of two such arrays, each having a length of 2>^ wave-lengths but
separated variously from to 2 wave-lengths in steps as noted. These
diagrams, of course, do not take into consideration the reaction re-
sulting from proximity to an antenna mast, located in such an opening.
The most important result is to emphasize the spurious lobes, as the
spacing between arrays is increased.
A second effect of grouping which is of considerable interest is that
of varying the direction of transmission by altering the respective
phases betw^een two or more arrays or between sections of the same
array. In Fig. 14 a series of diagrams Is shown for a typical case of
two 3J4 wave-length arrays, spaced one wave-length. All elements in
the same array are driven in phase, but the two arrays differ in phase
by various amounts, as noted. It will be observed that the possible
rotational effect is very limited. The general equation for this diagram
is given by formula (36) of the mathematical appendix.
This effect was investigated further by assuming a continuous array
7}^ wave-lengths long, made up of 16 couplets spaced at intervals
of ^ wave-length. The results are depicted in Fig. 15. The top row
assumes that the array is divided into two sections of eight couplets
each. This gives similar but not exactly the same results as those of
Fig. 14. The array, however, might have been divided into other sec-
tions for purposes of phasing. The various possible combinations are
tabulated below:
Number of Number of Couplets
Sections per Section
2 8
4 4
8 2
16 1
Diagrams in rows two, three, and four show that, as the array
continues to be divided into smaller sections, the direction of trans-
mission is capable of greater variation without sensible loss of sharp-
ness. If the array be divided into two sections this range is limited
to perhaps 3 deg. as in the case depicted in Fig. 14. Although this is
very moderate, it is extremely useful in correcting for any errors in
the orientation of the supporting structure or possibly correcting for
deviation of the projected radiation caused by peculiarities of the ad-
jacent terrain.
78
BELL SYSTEM TECHNICAL JOURNAL
o o
O uj
-â– 5
.
M
:^
2 Z
II
L. <"
o ^
<
< 5
q u <
< a ^
II II
< -o
U
0. _i 5 rt
O uj I- ^
< < IE —;
O o <
t=|CM
GAIN OF DIRECTIVE ANTENNAS
79
t=|aj
bo
Qu.Q.
S03
UkJ
<<|CM
80 BELL SYSTEM TECHNICAL JOURNAL
If the array is divided into four sections the rotation may extend
over a range of perhaps 9 deg., while for eight sections it may be 15 deg.
The final case of 16 sections of one couplet each permits of considerable
flexibility such as would be useful in operating with several distant
stations in the same general direction. It should be pointed out, how-
ever, that the problem of making 16 phase adjustments each time a
station wishes to change its direction of transmission is of considerable
magnitude. For the particular case illustrated above it appears that
the maximum rotation of the projected radiation is more or less pro-
portional to the number of sections into which the array is divided.
It may readily be seen from the two top rows of diagrams in Fig. 15
that continued addition of phasing amounts effectively to negative
rotation. This may also be seen from an analysis of the equation of the
diagram.
Fields of Linear Arrays
The successful use of an array of couplets to give unidirectivity
suggests that the use of more than two parallel linear arrays might
further be employed to advantage.^ Obviously many such combina-
tions are possible, but one of some interest has been investigated
below. As a concrete example of this variation of gain with arrange-
ment of arrays, a series of diagrams for 36 elements has been plotted
in Fig. 16. The condition of spacing and phase intervals between
columns of each of 34X has been chosen. The horizontal character-
istic is given for separations between rows of both }4 and 34 wave-
length. The vertical characteristic common to these two separations
is also shown. The equation of the diagram is given in formula (17)
of the mathematical appendix below.
It will be observed from Fig. 16 that the horizontal directivity is
for the most part only moderate, but approaches a maximum for the
condition where a long broadside array prevails, whereas the vertical
directivity is increased by increasing the number of columns in the
field. A substantial loop will be found near the rear of diagrams corre-
sponding to an odd number of columns. It is of further interest that,
as far as horizontal directivity alone is concerned, the optimum may
be derived either from a single array of 36 elements or from 18 couplets.
Considerations of both minimum interference and total gain, however,
make the latter preferable. These conclusions may also be reached by
more direct analysis.'^
« U. S. Patent 1,643,323, John Stone Stone, September 27, 1927.
' Wilmotte, "General considerations of the directivity of beam systems," Jour.
I. E. E., 66, 955.
= RATH
M
â– REL/(
— 1
UATIOI
_ SIN
2 SI
-4
lUATlO
_ SIN I
4 SI
lUATia
8S|-
XUATI^
SIN
'^G
Re RATIO OF $
SUBORDINATE LOOPS '
BETWEE
TWO GROUPS OF EIGHT COUPLETS EACH
^ S,N2T,C^3IN<H.B7 . ^"'ty"'»' .COsJ^(l-COS»)
a SIN TT (4 SIN » + BT 8 SIN (f SIN *)
FOUR GROUPS OF FOUR COUPLETS EACH
. SIN a TI B SIN â– ^ B') SIN (2 II SIN t ) ^
DIRECTION OF
TRANSMISSION
EIGHT GROUPS OF TWO COUPLETS EACH -
. SIN STl (SIN 0-fB') SIN [TI SIN )
8 SIN n (2 SIN ^ + B*) 2 SIN (^ SIN ^)
SIXTEEN GROUPS OF ONE COUPLET EACH -
GOUATION OF DIAGRAM
16 TT (4 SIN a -4- B'l _
-cosf)
SIN 16 TT (^ SIN » 1- bQ
16SINTl(isiN« + B')'
n-O.II R-0.15
Fig. 15 — Effect of phasing between sections of an array.
ff— ^5in 90'
S'.tO
R=1.0
ARRANGEMENT
n COLUMNS
^ AT.
NOTES
^^k
1 TO THAT OF DIRECTIONAL DIAGRAM
ICLES TO THAT OF DIRECTIONAL DIAGRAM
I LOOPS TO THAT OF MAIN LOOP
ACING BETWEEN ELEMENTS IN SAME COLUMN
R»I.O
ARRANGEMENT
n COLUMNS
AX
.- NOTES
S'S.Z
R=.0O8
-^^^
TO THAT OF DIRECTIONAL DIAGRAM
CLES TO THAT OF DIRECTIONAL DIAGRAM
• LOOPS TO THAT OF MAIN LOOP
'ACING BETWEEN ELEMENTS IN SAME COLUMN
(HORIZONTAL PUtNE)
(VERTICAL PLA^e)
ARRANGEMENT OF ARR
^^^
•EQUATIONS OF DIAGRAMS -
SIN (n i [COS « - I])
n SIN (5 [cos «-0)
SIN (n; [SIN 6-0)
nsiN(I[s,Ne-0)-
LMC^n^t^.
SIN (N It A SIN
»)
N SIN 01 A SIN
»)
SIN (N IT A SIN
8)
N SIN (IT A SIN
8)
SIN (N TI A 5IN
0)
ns,N(5 [SINO+I])
Fig. 16 — Directional diagrams due to a field of thirty-s
> = RATIO OF AREA OF UNIT CIRCLE TO THAT OF DIRECTIONAL DIAGRAM
>'= RATIO OF AREA OF TANGENT CIRCLES TO THAT OF DIRECTIONAL DIAGRAM
* = RATIO OF AREA OF SUBORDINATE LOOPS TO THAT OF MAIN LOOP
V= FRACTION OF WAVE LENGTH SPACING BETWEEN ELEMENTS IN SAME C
ARRANGEMENT OF ARRAY —
DIRECTION
ISSION
NOTES
S = RATIO OF AREA OF ONE TANGENT CIRCLE TO THAT OF DIRECTIONAL DIAGRAM
R= RATIO OF AREA OF SUBORDINATE LOOPS TO THAT OF MAIN LOOP
OF WAVE LENGTH SPACING BETWEEN ELEMENTS
-EQUATIONS OF DIAGRAM
Fig. 17 — ^Vertical plane diagrams due to couplets of coaxial antennas — number of couplets versus separation in wave-lengths.
GAIN OF DIRECTIVE ANTENNAS 81
Stacked Antennas
Thus far the discussion has centered mainly around directivity
produced by placing vertical antennas in horizontal array. Added
gain may be had also by incorporating directivity in a vertical plane.^
This is frequently accomplished by arranging individual antennas one
above another with their axes coUinear, and is sometimes known
as stacking. The fundamental principles of analysis are the same as
those already utilized. However, an approximate correction must be
allowed to account for the fact that the radiation from a linear oscilla-
tor increases from zero along the axis to a maximum in a plane per-
pendicular to the axis. The directional characteristic in planes passed
through and parallel to such a radiator is approximated by two
tangent circles.
Fig. 17 shows a series of directional diagrams indicating the re-
sults of stacking unidirectional couplets. The diagrams shown refer
to the plane passed through the axes of the two linear oscillators com-
prising the couplet. On each diagram is a unit circle corresponding to
a single point source. Inscribed are the two tangent circles, represent-
ing the vertical directional characteristic of a single linear source.
Inside one of the tangent circles is the final directional diagram of the
stacked array. The ratio of the area of the tangent circles to that of
the characteristic diagram is given under each figure. This may be
regarded as a rough measure of the relative gain. These diagrams are
arranged horizontally in order of increasing number of couplets and
vertically in order of separation. It frequently happens in practice
that each radiator is approximately >^ wave-length long so it is con-
venient to utilize a vertical spacing interval also of }4 wave-length.
Consequently the second row of diagrams is probably of greatest
practical interest. In calculating these diagrams earth efi"ects have
been ignored.
In Figs. 18 and 19, the gain in decibels to be expected from stacking
couplets has been plotted against number of couplets and fractional
wave-length spacing. These values, like those for Figs. 7 and 8 above,
were calculated by integrating the equation of diagram over a sphere
of arbitrary radius. This was accomplished by use of equation (30)
below. On account of the limited data at hand, Figs. 18 and 19
should be regarded only as a convenient method of illustrating the
trend of the variables. These indicate that somewhat lower corre-
sponding improvements result from stacking than from increasing the
length of an array.
8 U. S. Patent 1,683,739, John Stone Stone, September 11, 1928.
GAIN OF DIRECTIVE ANTENNAS 81
Stacked Antennas
Thus far the discussion has centered mainly around directivity
produced by placing vertical antennas in horizontal array. Added
gain may be had also by incorporating directivity in a vertical plane.^
This is frequently accomplished by arranging individual antennas one
above another with their axes collinear, and is sometimes known
as stacking. The fundamental principles of analysis are the same as
those already utilized. However, an approximate correction must be
allowed to account for the fact that the radiation from a linear oscilla-
tor increases from zero along the axis to a maximum in a plane per-
pendicular to the axis. The directional characteristic in planes passed
through and parallel to such a radiator is approximated by two
tangent circles.
Fig. 17 shows a series of directional diagrams indicating the re-
sults of stacking unidirectional couplets. The diagrams shown refer
to the plane passed through the axes of the two linear oscillators com-
prising the couplet. On each diagram is a unit circle corresponding to
a single point source. Inscribed are the two tangent circles, represent-
ing the vertical directional characteristic of a single linear source.
Inside one of the tangent circles is the final directional diagram of the
stacked array. The ratio of the area of the tangent circles to that of
the characteristic diagram is given under each figure. This may be
regarded as a rough measure of the relative gain. These diagrams are
arranged horizontally in order of increasing number of couplets and
vertically in order of separation. It frequently happens in practice
that each radiator is approximately K wave-length long so it is con-
venient to utilize a vertical spacing interval also of ^2 wave-length.
Consequently the second row of diagrams is probably of greatest
practical interest. In calculating these diagrams earth effects have
been ignored.
In Figs. 18 and 19, the gain in decibels to be expected from stacking
couplets has been plotted against number of couplets and fractional
wave-length spacing. These values, like those for Figs. 7 and 8 above,
were calculated by integrating the equation of diagram over a sphere
of arbitrary radius. This was accomplished by use of equation (30)
below. On account of the limited data at hand, Figs. 18 and 19
should be regarded only as a convenient method of illustrating the
trend of the variables. These indicate that somewhat lower corre-
sponding improvements result from stacking than from increasing the
length of an array.
8 U. S. Patent 1,683,739, John Stone Stone, September 11, 1928.
82
BELL SYSTEM TECIIXICAL JOURNAL
13
12
I I
10
9
in
id «
^ 7
u
Q
Z 6
Z
< ^
o
4
3
2
I
>,J'
12.
4 >â–
2 3 4 5 6 7
NUMBER OF COUPLETS
Fig. 18 — Calculated gains from stacked antennas.
12
^-'''
y
,.^-'
.--'
10
9
8
7
6
5
'2^'
y
^'
r-"
r
^
..--'
^-'
-
^^ , .
--â–
---'
8^
y^
^â– ^
^
^-'
^ '
1
r
y
^^â– ^
^>
-'^
,--
-'
y
4^
^-'
''
>
''"
.-
-4
<'
3 ,
^^'
• ''''
^^-
'-'
""
'
2
,-'
''
3
'"
4 03 0.6 0.7 0&
FRACTIONAL WAVELENGTH SPACING
Fig. 19 — Calculated gains from stacked antennas.
GAIN OF DIRECTIVE ANTENNAS
83
Arrays Incorporating Both Horizontal and
Vertical Directivity
The gains of arrays combining both horizontal and vertical direc-
tivity may not be simply calculated by adding the gains (expressed
in decibels) corresponding to elements arranged respectively along the
two principal coordinate axes. However, they may be calculated
except for earth effects by means of equation (26) below. Some cal-
culations of this kind have been made and the data are tabulated below.
They assume a total of 36 couplets which are arranged variously as
noted. In the first case all 36 couplets are arranged as a simple
horizontal array. The second case assumes that they are arranged in a
TABLE II
Number of Couplets
Along Horizontal
Axis
Number of Couplets
Along Vertical
Axis
Gain over Single
Half- Wave Element
Decibels
N
N
G
36
1
19.7
18
2
19.0
12
3
18.9
9
4
18.8
6
6
18.7
4
9
18.6
1
36
17.5
Fig. 20 — Approximate three-dimensional diagram. Linear antenna array
reflector. Aperture two wave-lengths by eight wave-lengths.
with
broadside rectangle two elements high and 18 elements wide. This
combination may be regarded as two arrays of 18 couplets arranged one
above the other. The third case similarly assumes three arrays of
12 couplets each. A separation between couplets of j4 wave-length
has been assumed throughout. The most economical arrangement of
such an array depends not only on the relative costs of real estate
and towers but also on feed-line losses and effects due to the proximity
84
BELL SYSTEM TECHNICAL JOURNAL
of the earth. The latter have specifically been omitted in this dis-
cussion.
Fig. 20 shows roughly the calculated directional characteristics of
a typical stacked array incorporating both horizontal and vertical
directivity. The planes passed through the diagram serve only as
convenient references to assist in visualizing the horizontal and vertical
diagrams. Earth effects of course, have been ignored.
Appendix
A general case of linear arrays which includes those used exten-
sively in short-wave radio work, consists of a number of sources equi-
spaced and equiphased along each of the three principal coordinate
axes such that the space between sources is made up of rectangular
parallelopipeds with the individual sources located at each corner.
This may be regarded as A^ parallel planes each made up of N parallel
columns where each column is made up of n individual radiating ele-
ments. The arrangement is made more evident by Fig. 21. The
Fig. 21— General case of linear antenna arrays.
usual conventions for representing three-dimensional space have been
adopted. We may designate the spacing between elements along the
X, y, and z axes, respectively, by a\, A\, and A\ and their corresponding
phase displacements between adjacent elements along the three princi-
pal axes by bT, BT and BT.
The distance from any point in space to a particular radiator is
i?„jv^ = R - (N - l)A\ cos 6 (1)
— {N — 1)AX cos (p sin 6 — {n — l)aX sin 6 sin 0.
GAIN OF DIRECTIVE ANTENNAS 85
Similarly the time phase of any particular element relative to the
origin is
8nNN = Lin - l)b + (N - \)B + (N - l)5]r. (2)
The instantaneous value of the electric field at any remote point P
due to one of these sources is given by
It
En' = A cos — (C/ — Rn') -^ 8n' = A COS \l/n', (3)
where n' = nN.
The resultant interfering effect at a point P due to n' such sources
all of equal amplitude is given by
£2 = „'£o2 + 2£o-[cos (^/'x- 1^2)+ cos (^1- 1^3)+ cos (4^i-rPd + -- • etc.
+ cos (\l/2 - h) + COS (\p2 - h) + COS (l/'2 - V^s) H CtC.
+ COS (1^3 — ^i) + COS {\p3 — ib)-\ etc.
+ COS {rPn'-i-Ml. (4)
The summation above gives rise to three series as follows:
•S'l = (w — 1) cos 27r(a sin 6 • sin (j) -\- b)
+ (w — 2) cos 2-2x(a sin 6 - sin + 6)
+ (w — 3) cos 3-27r(a sin d • sin cf) -{- b) + • • •
+ cos (w — l)-27r(fl sin d • sin 4> -\- b), (5)
Sy = (N - 1) cos 2ir{A sin O- cos </> + 5)
+ (iV - 2) cos 2-27r(^ sin 6 - cos + 5)
+ (N - 3)cos3-2Tr{As'md • cos + 5) + • • •
+ cos (N - 1)-2t{A sin 6 - cos + B), (6)
S, = {N - 1) cos IwiA cos 6 + B)
+ (// - 2) cos 2-2t(A cos + 5)
+ (iV - 3)cos3-27r(^cos0 + 5) + •••
+ cos (A^ - 1) •2Tr{A cos 9 -{- B), (7)
such that
£2 = £o2(^ _j_ 25,) (A^ + 25,) (A^ + 2S,). (8)
Each series is of the type
S = (ti — 1) cos X -\- (n — 2) cos 2x
+ (» - 3) cos 3;c + • • • + cos (w - l)x-' (9)
86 BELL SYSTEM TECHNICAL JOURNAL
which is readily summed giving
so
, ^ ^ (cos nx - 1)
(cos X — I)
sm^y
sm^-
sin mr{a cos 4> - sin + &)
sin iria cos ^ • sin ^ + 6)
sin NiriA sin </> • sin + 5) sin
NiriA cos + 5)
sin ir(^ sin • sin ^ + 5) sin
7r(yl COS d -\- B)
(10)
(11)
Reducing to common voltage level and including a term sin d to cover
the case of radiation from Hnear oscillators we have for the equation
of the directional diagram
_ sin mr{a cos <^ • sin + ^)
n sin 7r(a cos ^ • sin + 6)
sin Nir{A sin (/> • sin g + .B) sin iVx(i4 cos 6 -{- B) ^.^ ^ .^2)
* iVsin Tr{A sin </> • sin + 5) ' iVsin t:{A cos 6 + B) '
It will be recognized that this equation is made up of four factors.
The first three account for the effects of the disposition of elements
along the x, y, and z axes, respectively, while the fourth, of course,
accounts for the direction of radiation from a linear oscillator. This
is an equation giving magnitudes only. In plotting polar diagrams
from this equation negative signs have no physical significance, and are
plotted in a positive sense.
An examination of this equation shows that there are many possi-
bilities which allow radiation in preferred directions, and at the same
time limit it in others. Some of these are discussed below.
Special Cases
If we assume n = 2, a = I, b = — j and B = B =
sin (NttA sin <^ • sin 6) sin (NttA cos d)
N sin {tA sin </> • sin d) N sin {ttA cos 6)
• cos- (cos • sin — 1)- sin 6. (13)
This corresponds to the practical case of transmission along the
x axis from an antenna curtain and reflector made up of N vertical
columns of N elements each.
GAIN OF DIRECTIVE ANTENNAS 87
The equation for the diagram in the (XY) plane may be had by
placing 6 — ir/2 giving
sin (iVx^ sin </)) tt, ^ .. ,. ..
^ = AT • / — 1—- — ttcos- (cos 4> — 1), (14)
iVsm (tt^I sm ^) 4^ ^ ^
which is the equation of the diagrams in Fig. 5 above. The corre-
sponding equation for the principal vertical section may be had by
placing = and = tt giving
sin (NttA cos 6) tt , . . <n • „
cos — (sm 6—1) sm d
Nsin {tA cos 6) 4
and
sm (NtA cos d) T . a \ A\ • a
-rr^—. 7 -. 77 COS — (sm 0+1) SHl Q
Nsm (ttAcos 6) 4
(15)
which is the equation for the diagrams of Fig. 17.
The diagram of a single linear array of point sources is specified
by the first term of equation (12) where 6 = x/2 or
sin W7r(a cos <^ + 6) ,...
^~ = 7 , I , X • (16)
n sm 7r(a cos <p -\- o)
The diagrams of Figs. 3 and 4 above may be calculated from equation
(16; by placing w = 2 and n = 16, respectively. This also agrees with
Foster's equation (1), page 307.^
The diagram of a field of coplanar linear arrays such as depicted
in Fig. 16 above follows from equation (12) by placing N = 1, a = I
b = - I SLXid B = 0.
If the diagram is to be restricted to the (XY) plane, 6 = t/2 and
• /TVT A • ,\ sin I w- (cos â €” 1) I
_ sm (NtA sm 0) ^ \ 4 ^ V ^ . ^.
^ ~ iV sin (tt^ sin 0) ' . / tt , ^ ,,\' ^'^
w sm I - (cos (/) — 1) I
Calculated Gains from Arrays
The flow of power through each unit area due to an advancing
electric wave is given by the Poynting vector as
s=^EXH, (18)
47r
where E and H are vectors representing respectively, the electric and
magnetic components of the advancing wave.
2 Loc. cit.
88 BELL SYSTEM TECHNICAL JOURNAL
For free space \E\ = |//| so
s=^E^. (19)
Now the total power radiated through a sphere enclosing an array
of sources is
Pj = Csda =^ r r^ E,"^ sin ed4>dd. (20)
A second system would give
t-Jo Jo
p. =^ j I £2' sin ed<i>dd. (21;
The radiated powers of these two systems might be so adjusted
at the source as to give equal fields at any point along a preferred
direction. A ratio of these powers, therefore, would be a convenient
measure of the relative directional properties of the two arrays. This
"test ratio" may conveniently be set up in terms of the equations of
the diagrams derived above. In which case
ri2 sin ed(i>d6
Jo Jo
^2" sm
If we assume all comparisons are to be made with respect to a single
linear oscillator the denominator reduces to 87r/3, so
r = A r f"" ri' sin ed(f>dd. (23)
>^Jo Jo
This ratio may conveniently be expressed in decibels. In which
case G = 10 logio l/T is sometimes called the gain of an array.
If we are interested in the solid array shown in Fig. 21, where
n-N'N linear oscillators, each having respective space and phase
separations of a\, bT; A\, BT\ and A\, BT, are arranged progressively
along the three principal coordinate axes, this becomes
_ 3 r^" r^' sin^ [t?7r(a cos </> sin + Z>)]
8^ Jo Jo ^^ ^^"^ ^'^^^ ^°^ (i> sin d -\- b)2
sin'' iNxjA sin sin g + Jg)]
* N^ sin2 liriA sin sin + 5)]
. ^f^^^^f/'^'+^ll ^sin^ed^de. (24)
N^ sm^ \_Tr{A cos 6 + B)j
CAIN OF DIRECTIVE ANTENNAS 89
This integration has been carried out by R. M. F'oster who has very
kindly placed the results at the writer's disposal. Only the final
result is given herewith:
T = -4Tr + -^/'S' (" - ^y cos {2irkb)'Q{2-Kka, 0)
+ -1^/e {N - K) • cos (2TKB)-Q(2TrKA, 0>
ni\ ly K=i
+ ^^^2 e| (N-K)- cos {2irKB) • (2(0. 2tKA)
+ -^"E e\« - ^)(iV - K)- cos (2x2^5)
11 I\ I\ i=l A'=l
cos (27rkb)-Q(2w^kW + XM2, 0)
K)(N - K)- cos (27rir5)
cos {2irKB)-Q{2TrKA, 2tKA)
,N - K)- cos (27ry^6)
cos {2TvKB)-Q{2Trka, 2TrKA)
+ -tItf/I:' e' (A^ - X)(iV - ^). cos {2^KB)
ni\^I\^ K=i K=i
+ -tI^oE' E' (« - ^)(iV - ^)- cos (2Tkb)
I 2 n-l x-1 A^-l
+ -TTT^2 L Z E (« - -^)(A^ - X)(A^ - iC)
• cos (2 7ry^6)- cos (I ttKB)- cos (2x^5)
• Q{2Uk'-a' + X2^2^ 2T;;:i4). (25)
Where the function
'2(-^'' 3') = (^2 _^ yj3/2 sin (V.x-2 + /) + ^^, ^ ^3^, cos (Vx2 + /)
^.2 2-1/2
- (^iiqr^lpsin (Vx2 + /). (25a)
In particular
and
sin a: , cos x sin .t
(2(x, 0) = — — + —-. -r- (25b)
„,_ , 2 cos X , 2 sin x ,r.- s
(2(0, .r) = - --^- + -— - • (2^c)
Special Cases
(1) If we assume n = 2, a = \, h = — \ and 5 = 5 = 0, the test
ratio is given by
90 BELL SYSTEM TECHNICAL JOURNAL
?'i = 2lViV + 2Wn'% ^^' ~ ^^ ' <3(2^^^, 0)
+ WX72'^' t, N - K){N - K)-Q(2TrKA, 2irKA). (26)
This, like equation (13), corresponds to the practical case of
transmission from an antenna curtain and reflector each made up
of N vertical columns of N elements, all driven in the same phase.
(2) If we assume that no stacking is involved, then N = 1 and we
have for the test ratio for N couplets
T2 = 27v+ W^Sl^^ ~ -^)-'3(27rX^, 0)
_ 1 _L ^ V/y V) r sin27ri^^
- 2"7V + 2N^h ^^^ ^^ [ 2tKA (2^^
, cos lirKA sin 2tKA
' (27rKA)^ {2TrKAf
This equation was used in the calculation of the data given in
Figs. 6, 7, and 8.
(3) If we wish to apply equation (25) to the case of a single array
of N linear oscillators driven in phase we have n = N = 1 and B = 0,
so
Ts =^ + 1, e'(A^ - K)-Q(2tKA, 0), (28)
which differs from equation (27) by a factor of two. This indicates
that an array of N equiphased linear couplets gives twice the field in
the preferred direction as received from JV equiphased linear elem.ents
radiating the same power.
(4) Applying equation (25) to the extremely simple case of one
couplet, n = 2, a = J, b = — j and N = N = I and
T, = h (29)
(5) We may calculate the test ratio for a single stack of linear
couplets (earth effects not considered) by placing iV = 1, » = 2, a = j
b = — J, and 5 = and get
^^ = 2lV + A^ %[ (N-K)- (2(0, 2.KA)
1 3 ^=1
COS (2TrKA) sin {2TrKA]
2N N^K^i 'I {2TrKAf {2TrKA)'
(30)
CAIN OF DIRECTIVE ANTENNAS 91
This equation was used in calculating the data given in Figs. 18 and 19.
(6) The test ratio for the case of the rectangular array of nN
elements discussed in connection with Fig. 16 may be calculated by
placing N=l,a = i, b=— I and B = 0. In which case
^« = 4r + -4r2''^' (^ - k)-q(2tKA, 0)
+ -4r e' L {n - k){N - K) • cos (^)
7l~I\~ K=l k=l \ ^ /
Q(^2r^^+KU\oy (31)
Areas of Directional Diagrams
In general, the areas of directional diagrams may be calculated
from their equations by the usual integration methods. The special
case of N couplets in horizontal array, such as used rather generally
in practice and shown in Fig. 5 above, is of sufficient importance to be
given here. The area of the diagram in the (XY) plane is
S = -^,\^ + 'z\n-K)' Jo{2tKA) ' cos ItKB 1
(32)
This equation was used in calculating the data given in Fig. 5.
The area of diagrams in the horizontal plane due to a single array
of N oscillators is given by the equation:
S = ^
^ N'
N
+ "Z {N - K) • Jo{2tKA) • cos 2tKB 1 .* (33)
K=l J
This differs from equation (32) by a factor of two and indicates that
regardless of whether the gain is reckoned by an integration over a
unit sphere or in terms of the area of the horizontal diagram the effect
of the reflector is to double the radiated field in the preferred direction.
Placing -tV = 1 in equation (32)
S = h (34)
This is analogous to equation (29) above.
* R. M. Foster, "Directive diagrams of antenna arrays," Bell Sys. Tech. Jour., 5,
307; 1926.
92 BELL SYSTEM TECHNICAL JOURNAL
Arrays of Arrays
Each element of a generalized linear array, such as shown in Fig. 21,
may be replaced by a generalized array, thereby producing an array
of arrays.^ It may be shown that the resultant is given by an array
factor, representing the characteristics of individual arrays, times
other factors representing the relative position of the individual arrays
in the array of arrays. A derivation analogous to that beginning on
page 22 results in the equation
_ sin n'lria' sin + b')
K = T
n' sin 7r(a' sin (/> + h')
sin N'r{A' ^\n + B') sin N'-k{A' sin </> + B')
' N' sin Tr{A' sin + iJ') ' N' sin Tr{A' sin + 5') '
(35)
where a'\, A'X and A'\ are the coordinate spacings between arrays and
b'T, B'T, and B'T are the corresponding phase intervals, and r repre-
sents the characteristics of one of the individual arrays. If each array
is of the type shown in Fig. 5, r is given by equation (14) above.
Placing n' = N' = 1 and N' = 2 also n = 2 and B = 0, the above
equation reduces to
sin A^'7r(yl'sin<^ + 5') sin 7V7r(^ sin 0) t ,, , .-,,
-K = -T77— : \ A , ■, ^,\ ' TT—- T—A — = : COS -r (1 — COS (b), (36)
iV'sm7r(yl'sm<^ + j5') iVsm 7r(^ sin0) 4 ^j^ \ j
which is that made use of in calculating the diagrams in Figs. 14 and 15.
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GAIN OF DIRECTIVE ANTENNAS 95
Uda S "On the wireless beam of short electric waves," Jour. I. E. E. (Japan)
' (Reprint No. 20), July, 1928. ,
Walmsley T., "Polar diagrams due to plane aerial reflector systems, Exp. Wireless,
5, 575-577; October, 1928. „„ „^. ,, .-
Wells N "Beam wireless telegraphy," El. Rev., 102, Part I, 898-902; May 23,
' 1928; 102, Part II, 940-943; June 1, 1928.
Wilmotte R. M., "General considerations of the directivity of beam systems.
Jour. I. E. E., 66, 955-961; September, 1928.
Wilmotte, R. M., "The nature of the field in the neighborhood of an antenna,
Jour. I. E. E., 66, 961-967; September, 1928.
Wilmotte, R. M. and McPetrie, J. S., "A theoretical investigation of the phase
relations in beam systems," Joiir. I. E. E., 66, 949-954; September, 1928.
Yagi, H., "Beam transmission of ultra-short waves," Proc. I. R. E., 16, 715-741;
"Emissions radiotelegraphiques dirigees," V Industrie Elect., 37, 341-346;
August 10, 1928; 37, 372-376; August 25, 1928.
1929
Beauvais G., "Les ondes electriques tres courtes (15 i 20 centimetres)," Rev. Gen'
de I'ElecL, 25, 393-394; March 16, 1929.
Bechmann, von R., "Berechnung der Strahlungsdiagramme von Antennenkombi-
nationen," Telefunken ZeiL, No. 53, 54-60; December, 1929.
Campbell, G. A., U. S. patent No. 1,738,522, "Electromagnetic wave signaling
system," December 10, 1929.
Chireix, H., "French beam system for short waves," Bull, de la Soc. Frang. Radto-
telegraphique, 3, 79; May, 1929.
Chireix H., "French system of directional aerials for transmission on short waves,
Exp. Wireless, 6, 235-244; May, 1929. .
Gresky, G., " Richtcharakteristiken von Antennenkombinationen deren einzelne
Elemente in Oberschwingungen erregt werden," Zeits. f. Hochf., 34, 132-
140; October, 1929; 34, 178-183; November, 1929.
Hahnemann, W., German patent No. 474,123, "Einnchtung zum gerichteten
Senden und Empfangen mittels elektrischer Wellen," March 27, 1929.
Koomans, N., French patent No. 660,639, "Antenne directive," February 19,
1929.
Mathieu, G. A., "The Marconi-Mathieu method of multiplex-signaling, Marcom
Rev., 7, 1; April, 1929. o • -^ ^- ^f^
Mesny R, "Les ondes dirigees et leurs applications. Revue Saenttfique, No. 19,
577-585; October 12, 1929. ,
Moser, W., "Versuche iiber Richtantennen bei kurzen Wellen, Zetts. f. Hoch}.,
34, 19-26; July, 1929. . „ ^ , •
Ostroumov G. A., "A directional untuned short-wave receiving antenna, 1 eleg. t.
Tele}, b. Prov., 10, 111, 1929. „
Palmer, L. S. and Honeyball, L. L. K., "The action of a reflecting antenna. Jour.
I. E. E., 67, 1045-1051; August, 1929.
Pistolkors, A., "Calculation of radiation resistance of antennae composed of perpen-
dicular oscillators," Teleg. i. Telef. b. Prov., 10, 33, 1929. ,- ..^ .,„
Pistolkors, A., "Radiation resistance of beam antennas," Proc. I. R. E., 17, 562-^79;
March, 1929. „ ^ , r i.
Sammer von F., "Die Wirkungsweise von Drahtreflektoren, Telefunken Zett.,
No. 53, 61-71; December, 1929. o ii ..
Stenzel, H., tjber die Richtcharakteristik von in einer Ebene angeordneten btrahlem,
E. N. T., 6, 165-181; May, 1929. j r^ u ^ •
Strutt M J O "Strahlung von Antennen unter dem Einfluss der Erbodeneigen-
'schaften," Ann. d. Physik, Series 5, 1, 721-750 and 751-772; April 6, 1929;
Series 5, 4, 1-16; January 18, 1930.
Villem MR "La liaison radiotelephonique Paris — Buenos Aires par ondes courtes
projetees," Bull, de la Soc. Frang. des Elect., 9, No. 98, 1107-1145; October,
1929.
Yagi, H., German patent No. 475,293, "Einrichtung zum Richtsenden oder Rich-
tempfangen," April 25, 1929.
Absolute Calibration of Condenser Transmitters
By L. J. SIVIAN
Several methods have been used or proposed for the calibration of the
Wente condenser transmitter. The methods falling under the two classifi-
cations conveniently designated "constant pressure" or "pressure"
calibration and "constant field" or "field" calibration are most useful and
amenable to measurement. Which of these two calibrations is more
significant depends on the particular use made of the transrnitter. In the
following pages the methods now used or proposed are reviewed and the
advantages or disadvantages of each from the standpoint of transmitter
application are discussed.
IN the original design of the Wente ^ transmitter the effective
diaphragm resonance was well above 10,000 c.p.s. The new design
(Western Electric No. 394-Type), developed by Wente, has an
effective resonance at approximately 5,000 c.p.s. It is about ten
times more sensitive (on a voltage-pressure basis), and more immune
from effects of humidity and of barometric changes. The important
external dimensions of the instrument are shown in Fig. lA .
The response of the transmitter is defined as the ratio of the electro-
motive force generated to the acoustic pressure acting on the trans-
BACKPLATE-^
Fig. L4 — Contour dimensions of No. 394-type condenser transmitter.
Fig. IB — Contour dimensions of condenser transmitter used for field calibration.
1 See bibliography.
96
ABSOLUTE CALIBIUTION OF CONDENSER TRANSMITTERS 97
mitter. That ratio [i?(/) = e/p'], as a function of frequency, gives the
caHbration. Where and how is the acoustic pressure to be measured?
This can be done in any one of a number of ways, all of which in
general lead to different calibrations. The two calibrations most
useful and amenable to measurement are when the pressure is uniform
over the diaphragm and measured at the diaphragm and when the
pressure is the pressure in a progressive plane wave, undistorted by the
transmitter or any other obstacles; when the electromotive force is
measured the distortion of the sound field must be due to the trans-
mitter alone.
It is convenient to designate the former as "constant pressure" or
"pressure" calibration, the latter as the "constant field" or "field"
calibration. In general the field calibration will depend on the angle
of wave incidence. Incidence normal to the diaphragm gives the
"normal field" calibration. Where no confusion can arise, "field"
calibration will be used to imply normal incidence. The pressure and
field calibrations tend to coincide when the transmitter dimensions are
small compared to the sound wave-length and when there are no
appreciable impedances between the diaphragm and the sound field in
front of it. Neither condition obtains for the No. 394-Type Trans-
mitter, except at very low frequencies.
Which of the two calibrations — "pressure" or "field" — is more
significant depends on the particular use made of the transmitter.
Thus in the receiver testing machine, where the sound is substantially
uniform throughout a small chamber closed by the transmitter
diaphragm and by the receiver under test, the pressure calibration is
important. When the transmitter is used to pick up sound in the
open air at a distance from the source, the field calibration applies.
For other cases, neither calibration is directly applicable, this being
discussed at the end of the paper.
Constant Pressure Calibrations
For the several methods available for constant pressure calibration,
the pressure may be applied either acoustically or electrically. In the
acoustical group are the following methods :
1. Thermophone.2
2. Pistonphone.^' 2
3. Resonating tube.^
4. Compensation methods.
a. Electrodynamic compensation for acoustic pressure.*
b. Electrostatic compensation for acoustic pressure.^
5. Membranephone.
98
BELL SYSTEM TECHNICAL JOURNAL
In the electrical group for pressure calibration are the following
methods:
6. The back electrode (backplate) serving as the driving electrode.^- *
7. An auxiliary third electrode driving the diaphragm.
Ail but two of the above methods have been described in detail in
the articles to which references have been given so that only brief
descriptions of the methods are given in the following paragraphs.
1. Tliermophone. — The alternating pressure generated in the chamber
of which diaphragm D (see Fig. 2) is one wall, is computed from the
physical constants of the thermophone T, and of the gas (hydrogen)
filling the chamber. A computation similar to that in reference ^ is
discussed in Appendix I and II. The difference is in the manner in
Fig. 2 — Thermophone method.
which the heat conductivity of the walls is taken into account. Also a
slight correction for the yielding of the diaphragm is introduced, which
was superfluous with the earlier, less sensitive model. An important
advantage of the thermophone method is that it is not necessary to
have the heating element parallel to the diaphragm. This makes it
applicable to transmitters with curved or corrugated diaphragms. In
such cases it is difficult to provide the accurately parallel and narrow
spacing between the diaphragm and driving or compensating electrode,
required in electrostatic methods.
2. Pistonphone. — The pressure is generated by means of a recipro-
cating motor-driven rigid piston as shown in Fig. 3. The piston
amplitude is computed from the dimensions and the angular velocity
of the cam driving it. The motor drive makes the method suitable for
relatively low frequencies, up to about 200 c.p.s.
ABSOLUTE CALIBRATION OF CONDENSER TRANSMITTERS 99
3. Resonating Tube. — The pressure at the diaphragm end of the
tube (see Fig. 4) is computed from a measurement of the air particle
velocity at a pressure node. That velocity is obtained by observing
the deflection of a Rayleigh disk, R. D., placed in the tube. The
sound source R is shown as a moving coil receiver.
4. Compensation Metlwds. — The pressure in the chamber is de-
termined by measuring the force required to prevent motion of a
=Q=
MOTOR
Fig. 3 — Pistonphone method.
small auxiliary diaphragm Di, Fig. 5. With the sound pressure so
determined the corresponding electromotive force of the transmitter is
measured. The rest condition of D2 is indicated by absence of sound
in an exploring tube communicating with the space back of Di or by
absence of frequency variation in a high frequency circuit in which D^.
is made one plate of a condenser controlling the oscillation frequency.
k\\\\\\\\\\\ \\\\v\\\\\\\\v\\\\\\v\\\\\\\\\\\\\\\\\\\\\\\v\\\vv^^s^?^^^^s^
X'
Fig. 4 — Resonating tube method.
4a. Electrodynamic Compensation for Acoustic Pressure. — The com-
pensating pressure is provided by sending a current of adjustable
frequency, amplitude and phase through D2 placed in a steady magnetic
field.
4&. Electrostatic Compensation for Acoustic Pressure. — The same end
is attained with a potential difference of adjustable frequency, ampli-
tude and phase applied between D^ and a fixed electrode parallel to it.
100
BELL SYSTEM TECHNICAL JOURNAL
In particular the transmitter diaphragm and backplate may serve as
D2 and the fixed electrode. This, however, requires caution. The air
gap is so small (approximately 2.5 X 10~^ cm.) that unavoidable
variations in its value will in general cause appreciable variations in the
value of the electric driving force over dififerent parts of the diaphragm.
The non-uniformity of the air gap is due to mechanical imperfections
and to the electrostatic pull of the polarizing voltage. Furthermore,
in transmitters of the type here considered, the backplate diameter is
substantially smaller than that of the diaphragm, and hence the
compensating electric force is not effective in a peripheral portion of
the diaphragm.
V k k ■. ^ ^ ^ ^ k k k k k t k k k k ^ ^ k ■.». u k ■■■.■. ^ ^1 /T?^ "I
Fig. 5 — Electrodynamic compensation method.
The electric force in this case is provided by inserting between the
diaphragm and the backplate a steady potential difference, Fo, and a
much smaller alternating potential difference, Vi sin wt, in series. One
of the resultant force components is aV^Vi sin o^t which has the same
frequency as the sound source (e.g. a thermophone). The amplitude
and phase of the electric force are adjusted until it balances the
acoustic pressure on the diaphragm. This gives the value of the
acoustic pressure, provided a is known. The compensating electric
force is then removed, and the output of the transmitter due to the
acoustic pressure is measured. Thus the pressure calibration is
obtained. The value of a is given by a measurement of the value of
Fo required to balance a known static gas pressure established at the
ABSOLUTE CALIBRATION OF CONDENSER TRANSMITTERS 101
face of the diaphragm. This must be done for each instrument to be
caUbrated.
5. Membranephone. — In principle this method is similar to the
pistonphone. An acoustically driven membrane M (see Fig. 6)
replaces the motor-driven piston. From the volume displacement,
AV, of M the pressure on the transmitter diaphragm D is computed.
The value of A V is given by a measurement of the alternating variation
in capacitance between M and an auxiliary perforated electrode G.
The range of the method is from the lowest frequencies up to those at
which the linear dimensions of the chamber become comparable with
the sound wave-length (X). As with the thermophone, that upper
limit can be extended through the use of hydrogen instead of air.
The computation of A F is given in Appendix III. It will be noted
that the computation is independent of the mode in which the mem-
brane vibrates. However, for frequencies above the first resonance of
TO AMPLIFIER-
RECTIFIER
Fig. 6 — Membranephone method.
the membrane the requirement as to smallness of chamber dimensions
relative to X, becomes much more stringent than in the thermophone
case.
Methods Employing Electrical Drive. — Since the driving forces in this
group are electric the pressure on the diaphragm is affected by the
acoustic load on the front face of the diaphragm. To obtain the true
pressure calibration that acoustic load must be known. Practically
this is taken care of by making that load sufficiently small, rather than
accurately determining its value.
6. The Back Electrode Serving as the Driving Electrode. — The alter-
nating potential difference, Vi sin w/, is impressed in series with the
steady potential Fo, see Fig. 7. This gives a driving force component
aFoFi sin wt. The corresponding alternating variation in the trans-
mitter capacitance is determined by having that capacitance control
the frequency of a high frequency oscillator circuit. Absolute values
are obtained by means of a static pressure calibration as in Method 4.
102
BELL SYSTEM TECHNICAL JOURNAL
In this case, however, that does not give the force acting on the dia-
phragm unless the air impedance between the diaphragm and back-
plate is negligible in comparison with that of the diaphragm itself.
Hence the method does not apply to the No. 394-Type Transmitter.
The same consideration as to non-uniformity of the driving force over
the area of the diaphragm which was mentioned in connection with
Method 46, applies to this case.
7. Auxiliary Third Electrode Driving the Diaphragm. — Here an
auxiliary electrode M and a circular metal screen furnishes the electro-
static drive (see Fig. 8). It has nearly the same diameter as D and is
parallel to it. The gap between M and D is about thirty times greater
'-<2r
TO HIGH FREQUENCY
OSCILLATOR
HIGH
FREQUENCY
CHOKE
^hKSH
Fig. 7 — Electrostatic method — Back electrode serving as driving electrode.
than between D and the backplate. Hence the electric force on D is
uniform over the surface of D, and its absolute value can be computed
with some accuracy. The calculation is given in Appendix IV. Care
must be taken to avoid acoustic loading of Z) in a manner that would
materially change its impedance. With this possibility guarded
against, this method admits of an absolute transmitter calibration
from 20 to 20,000 c.p.s. A comparison of a calibration so obtained
wdth that given by a thermophone for the same transmitter,* is shown
in Fig. 9. The two are quite independent. The discrepancy between
the two up to about 6,000 c.p.s. is regarded as being within limits of
experimental error. The acoustic load imposed on the diaphragm by
* This particular instrument happened to be about 4 db less efficient than tlie
aAcrage No. ,S94-T}pe Transmitter.
ABSOLUTE CALIBRATION OF CONDENSER TRANSMITTERS 103
the calibrating apparatus, while relatively small in either case, is not
the same for both methods. At higher frequencies other factors
contribute. At the highest frequencies, say above 10,000 c.p.s., the
pressure on the diaphragm probably is more uniform in the present
method than in Method 1.
Constant Field Calibration
For constant field calibration methods it is difficult to provide a
plane progressive wave over a sufficiently large wavefront. Instead a
small source in a chamber lined with highly absorbing material is used.
The resultant progressive spherical wave, at sufficient distance from
igSIN 2ujt
TO AMPLIFIER-
RECTIFIER
Fig. 8 — Electrostatic method — auxiliary third electrode driving diaphragm.
the source, gives approximately the desired sound field. The measur-
ing device must give the absolute value of the undistorted field in-
tensity. We shall not consider the thermal, optical and sound radi-
ation pressure methods possible, on account of the experimental
difficulty which they present. One other absolute method is more
readily available:
The Rayleigh Disc, which on certain assumptions gives the absolute
value of the particle velocity in the sound wave. In the sound field
presupposed for the field calibrations, the corresponding sound pressure
is easily computed.^
Another procedure is to measure the sound pressure with the aid of a
"search transmitter." This is a transmitter whose dimensions are so
104
BELL SYSTEM TECHNICAL JOURNAL
small relative to the sound wave-length that its pressure calibration, as
obtained say by Method 1, may be taken to coincide with its field
calibration.
The normal field calibration of a No. 394-Type Transmitter is
shown in Fig. 10. The contour of the particular instrument used is
shown in Fig. \B. It was suspended from a thin rod clamped to the
metal band B. The measurements were made with a Rayleigh disc
(0.5 cm. diameter, 2.46 second period), using the modulated sound
method.^ The transmitter was placed 32 cm. from the sound source, a
1-cm. diameter tube attached to a loud-speaking receiver. The data
obtained for frequencies below 500 c.p.s., are believed to be not so
reliable as the rest because of appreciable reflections from the chamber
walls.
A = THERMOPHONE CALIBRATION
B = ELECTROSTATIC CALIBRATION
n_e^, -0.05a MILLIVOLTS
*^~p '" BAR
20
500 1.000
FREQUENCY IN rvCLES PER SECOND
6,000 10.000 20,000
Fig. 9 — Comparison of two pressure calibration methods.
For purposes of comparison, the pressure calibration (Method 7) of
the same instrument is shown. At the lowest frequencies the two
calibrations nearly coincide, as might be expected. At high fre-
quencies, say from 1,000 c.p.s. upward, the divergence of the two is
quite marked. It has been pointed out by several writers that the
difference may be regarded as due to two effects. First, i" as X de-
creases, the transmitter tends to cause a doubling of the pressure in
front of it as would a rigid wall. Second," the recess in front of the
diaphragm (Fig. 1) introduces a broad resonance which has its maxi-
mum approximately at 3,500 c.p.s. An estimate of this effect is given
in Appendix V.
The observed differences between the field and pressure calibrations,
from 500 to 8,000 c.p.s. are in fair agreement with those computed for
ABSOLUTE CALIBRATION OF CONDENSER TRANSMITTERS 105
the two effects given above. The computations are based on as-
sumptions as to the transmitter contour which are quite removed from
the actual case. Thus for the first effect it has been suggested that the
transmitter may be replaced by an "equivalent" rigid sphere of equal
>
N
A
\
\
\
A= PRESSURE CALIBRATION
B= FIELD CALIBRATION — NORMAL INCIDENCE
n e loO.O^a MILLIVOLTS
"-p-'° BAR
1
Fig. 10.4-
50 100 500 1,000 5,000 10,000 20,000
FREQUENCY IN CYCLES PER SECOND
-Pressure and field calibrations of No. 394-type condenser transmitter.
y
,— .
N
"-
\
\
\
\
\
\
\
1
A= NORMAL INCIDENCE
B= RANDOM INCIDENCE
n-fi -io0.05a MILLIVOLTS
" p-'° BAR
\
\
Ip^b
20 50 100 500 1,000 5,000 10,000 20,000
FREQUENCY IN CYCLES PER SECOND
Fig. lOB — Field calibrations of No. 394-type condenser transmitter for normal and
random incidence.
volume ^2 or of equal diameter.^^ The data in Fig. 10^ are best fitted
by assuming a sphere of 9 cm. diameter, i.e., a diameter even larger
than that of the transmitter. For the second effect the assumption is
made that the face of the transmitter acts as an infinite wall, and that
106 BELL SYSTEM TECHNICAL JOURNAL
the air particles in the recess aperture all move in phase and normally
to the diaphragm.
At still higher frequencies the doubled pressure effect largely persists
and superposed on it are a number of rather complicated diffraction
effects. These involve radial wave propagation across the diaphragm
recess while the above two effects are due to normal plane waves.
The marked dip at 11,200 c.p.s. corresponds to a sound wave-length
such that
-^li^PQ)' + (PAY -PA=\
"2^
(see Fig. lA).
So far normal incidence of the sound wave has been assumed. For
other directions of arrival, substantially different field calibrations are
obtained. Since the transmitter is symmetrical about any diaphragm
diameter, the effect of direction may be given in terms of the azimuth
angle of incidence. A set of azimuth curves for various frequencies
are given in Fig. 11, all expressed relative to the normal field cali-
bration. In general, the higher the frequency the greater the effect of
azimuth. For a large range of angles that effect is as great as or greater
than the difference between the pressure and the normal field cali-
brations. It is interesting to note that the anomalous azimuth curve
at 11,200 c.p.s. corresponds to a pronounced dip at that frequency in
the normal field curve.
Relation of Field Calibration to Actual
Transmitter Performance
We now consider the bearing of field calibrations upon the response
of the No. 394-Type Transmitter under one or two conditions of
actual use.
First, consider the case of a person speaking directly toward the dia-
phragm. The normal field calibration approximately applies, provided
the distance is not great enough for reflected waves to be comparable
with the direct wave and the distance is not so small that the transmitter
reacts back on the source (the voice), or that pronounced standing
waves are set up between the transmitter and the head. Outdoors and
in a well damped room distances ranging say from 6 inches to 3 feet are
likely to be within the above limits for the important voice frequencies.
On the other hand, for much of indoor work the distances from the
microphone to the source and to the several reflecting surfaces are such
that waves reaching the microphone by reflections are comparable
with and often predominate over the direct sound. Besides, the
microphone often is so placed that the direct sound strikes it more
ABSOLUTE CALIBRATION OF CONDENSER TRANSMITTERS 107
nearly at a 45-degree or 60-degree angle rather than normally. In a
29-foot X 29-foot X 13-foot room having a reverberation time of 1
180
SOURCE 32.0CM.
FROM DIAPHRAGM CENTER
Fig. 11 — Azimuth response of No. 394-type condenser transmitter.
108 BELL SYSTEM TECHNICAL JOURNAL
second, the reflected waves reaching the microphone at 12 feet from a
small source contribute much more to the microphone output than
does the direct sound. To illustrate the effect of these reflections, the
curve {b) in Fig. 105 has been plotted. It is based on the data of Fig.
10 and Fig. 11, and assumes that the transmitter is acted upon by
progressive plane waves arriving with equal intensity from all directions
in space. Their phases are taken to have random distribution. At
any one frequency the response of the transmitter is then proportional
to
41
U
lA{d)J â– sin e-dd,
where A{^) is the azimuth factor taken from Fig. 11. The result is
seen to be intermediate between the pressure and the normal field
calibrations, for frequencies up to about 8,000 c.p.s. Under these
circumstances it is immaterial which way the diaphragm faces, but
this holds only for sustained sounds. For sounds of short duration,
the peak amplitudes in the microphone output often are of particular
interest. They will be more nearly given by that single field curve
corresponding to the azimuth with respect to the sound source in which
the transmitter happens to be.
The above discussion of directional effects is simplified by the fact
that the No. 394-Type Transmitter is symmetrical about any dia-
phragm diameter. Hence a single parameter — azimuth angle — is
sufficient. The amplifier mounting cases usually employed destroy
that symmetry. The directional effect becomes much more compli-
cated since it involves two parameters, e.g. two direction cosines of the
diaphragm axis. It has been suggested ^^ that this complication can
be done away with by placing the transmitter and its amplifier case in
a rigid hollow sphere, only the transmitter front being exposed. If
the front contour of the instrument be designed closely to conform to
the rest of the sphere, and if the diaphragm subtend a sufficiently
small angle at the center of the sphere, the directional effect can be
computed. ^^
The simplest directional properties, i.e. uniform response for all
directions of incidence, require a transmitter whose linear dimensions
are small (say < }iX) relative to the shortest sound wave-length to be
picked up. For a frequency range extending to 10,000 c.p.s., this
means a transmitter less than 0.85 cm. in diameter. In general such
restriction on the permissible size adds to the difficulties of construction
and operation of the instrument. It is not intended to imply that non-
ABSOLUTE CALIBRATION OF CONDENSER TRANSMITTERS 109
directivity of the transmitter is always desirable for pickup systems of
highest quality.
A complete description of the performance of the microphone as an
electro-acoustic converter is extremely complex. It involves the
microphone, the sound source, their relative positions, and the sur-
rounding acoustic configuration. Furthermore, it is limited to sound
sustained long enough to allow the reflection pattern to attain a steady
state. Therefore, in order to obtain a reasonably simple and useful
statement of the transmitter response, the field calibration is made
under the ideal acoustic conditions stated in part A. Even then the
field calibration (including, of course, the azimuth measurements) is far
more difficult and laborious than the corresponding pressure cali-
bration. For some important purposes the pressure calibration is
sufficient, even though the transmitter be intended for use in an
"open" sound field. An instance is the specification and comparison
of instruments having similar contours. The difference between the
field and pressure calibrations, once determined for an individual
instrument, applies to all others. That is, provided the acoustic
impedances of their diaphragms are not too widely different, which
usually is the case. Therefore the response of any instrument, as a
function of frequency, age, barometer pressure, temperature, etc., is
given by the pressure calibration. The thermophone method (Method
1) is particularly suitable for rapid and reproducible determinations of
the pressure calibration. That is the method employed for the
specification of No. 394-Type Transmitters, and of others having
similar contours, in the Master Reference Systems ^^ for Telephone
Transmission in Europe and in this country.
I am indebted to Messrs. R. T. Jenkins, H. T. O'Neil and E. M.
Little of Bell Telephone Laboratories for much of the material used in
this paper.
Appendix I
The pressure generated by the thermophone is slightly reduced by
the heat conductivity of the chamber walls. That conductivity is so
great as compared with that of the gas, that zero temperature variation
at the walls may be taken as one of the boundary conditions of the
problem. This results in a solution nearly identical with that of eq.
(7), p. 336, in the original derivation.^ The correction factor given
there on p. 340, which takes care of the wall conductivity, is now
found to be more nearly unity. The difference between the two
solutions is shown in Fig. 12 for a special case typical of condenser
transmitter calibrations. As might be expected, it is greater the lower
the frequency.
no
BELL SYSTEM TECHNICAL JOURNAL
^
^
^
\
N
S
s
^
^
-
^— 5— -
^
^
"
'
.^"^
s
s
V
^
s
s
^«
A=NEW FORMULA
B=OLD FORMULA
C = DIFFERENCE BETWEEN A AND B
- D inOOSa BARS
s
P^^ = l00.05a
Pb=Pa-ioOO-
VOLTS AC ACROSS THERMOPHONE
BARS
VOLTS AC ACROSS THERMOPHONE
v
N
\
\
s
\
500 1,000
FREQUENCY IN CYCLES PER SECOND
5,000 10,000 20,000
Fig. 12 — Pressure generated by a thermophone in a transmitter calibration chamber.
Appendix II
In the thermophone theory the walls of the chamber were treated as
being rigid. Actually the transmitter diaphragm presents a small but
finite admittance in shunt with the elastic admittance of the gas in the
chamber. The correction factor M due to this, is approximately
M =
1
l+(2M!+2^.cos
Vo
Vo
where
7 = 1.4=^,
assuming adiabatic conditions
po = 10^ bars atmospheric pressure,
Vo = volume of thermophone chamber,
V = volume displacement of diaphragm per bar,
6 = phase angle of above displacement with respect to the pressure on
the diaphragm.
At low frequencies cos 6 may be taken as nearly unity, and v can be
approximately computed as below
ABSOLUTE CALIBRATION OF CONDENSER TR.ANS HITTERS 111
1 ai^ yi
AC ,j s
2a.i2 h - 71
2ai2V "^ // / '^ 3ai' h -
AC C3 ^1 ,
2gi2
where // = separation between diaphragm and back plate without
polarizing voltage.
Ci = capacity between diaphragm and back plate without
polarizing voltage.
C2 = above capacity in presence of polarizing voltage.
Cz = total transmitter capacity, with polarizing voltage.
£0 = polarizing voltage.
ei = transmitter e.m.f. per bar, uncorrected for yielding of
diaphragm.
a I = diaphragm radius; 02 = back plate radius.
For the 394-Type Transmitter, up to about 2,500 c.p.s., .1/ is nearly
0.92. Above that the correction decreases owing to decreasing cos 6,
and becomes negligible at 5,000 c.p.s. For still higher frequencies the
correction becomes negative but remains small due to the increasing
diaphragm impedance.
Appendix III
Schematically the membrane phone is shown in Fig. 6. D is the
diaphragm of the transmitter to be calibrated ; M, a stretched membrane
acoustically driven from the receiver R; G, a. perforated plate. Let
V = volume between D and M; yo = normal separation between G
and AI; Co = normal capacitance between G and M.
Then, if yoll + KiS) • sin co/] represents the GAI separation when M
is driven by R, the resultant capacitance variation is:
K(S)
z\c = sm cot- -—J — • I
41lyo J 1 +
and
AC = sin CJt- -7-T — • I :; — ; — j^. . ... ■", 'dS
AV = sin cot-yo I K{S)-dS,
the integration extending over the entire area of M.
112 BELL SYSTEM TECHNICAL JOURNAL
Taking K{S) < < 1, but without restrictions on the variation of
K{S) over the surface of M,
AV = 4n3/o^-ACo.
Hence the transmitter sensitivity is given by
62 e^VEo . .,
R = — = 3^ volts/bar.
The above presupposes: (1) V/S<<X, ^jS<<X•, (2) acoustic
admittance of D is very small compared with that of V; (3) adiabatic
compression. If necessary, corrections for deviations from (2) can
be made in accordance with Appendix II. The correction for (3) is
found to reduce the pressure in the ratio
R' = R ^
l + (^_l),tanh^a'
where
13a
;5 = (. + i)V#.
when C = specific heat at instant pressure,
K = thermal conductivity of the gas,
p = density.
The upper frequency limit imposed by condition (1) can be raised by
filling F with hydrogen. For the No. 394-Type Transmitter, and with
R a No. 555-W Western Electric Receiver, an air-gap yo = 0.075 cm.
corresponds to easily measurable values of ei and e^. M was a 0.001
inch duralumin diaphragm, stretched to 5,000 c.p.s. resonance fre-
quency. It was found that the upper frequency limit of the method is
determined by M breaking up when vibrating in one of its higher
natural modes. This tends to produce a non-uniform pressure on D,
and the above condition must be met much more perfectly than in the
thermophone case.
Appendix IV
The particular electrostatic calibration described below, employs a
separate driving electrode and a sinusoidal driving voltage which
produces a sinusoidal driving force of double frequency. The latter
has the advantage of adding frequency selectivity to shielding as the
ABSOLUTE CALIBRATION OF CONDENSER TRANSMITTERS 113
means for keeping the relatively large driving voltage out of the
transmitter output circuit.
In terms of Fig. 8 the sensitivity of the transmitter is given by
i?=ivol,s/bar, . = ,..„.10-»«-. P = ,^,^^^,^.j,,. .
where V^!2 = Vi, measured in volts, iaV2 = ii, h = separation
between AI and D. The e.m.f. e is measured by means of the potential
attenuator P, carrying a known current ia, and having an input
resistance /-„. At any one frequency two quantities must be measured:
V, say with an electrostatic voltmeter, and a, the setting of the
attenuator in decibels. The current ia must be known but can readily
be kept constant at all frequencies if a heterodyne oscillator be used as
the source.
Two corrections must be applied. First, the auxiliary electrode is
perforated. Hence not all of its area is electrostatically effective.
Second, p in the above is the electrostatic force per unit area, rather
than the acoustic pressure on the diaphragm. The two are different,
in general, because of the acoustic load (Zd) on the front face of the
diaphragm. The value of Zd is affected by form of C, the auxiliary
electrode, and by the acoustic impedance beyond it in the chamber G.
It is best to have Zd as small and as free from reactance as possible.
This is accomplished by using stretched fine metal gauze. Copper
gauze, 300-inch mesh, is quite good. It terminates in the tube TT,
3>^-inch iron wall, which is filled with several layers of loose cotton
batting and hairfelt. The effectiveness of the arrangement was
judged by the fact that altering the size of G did not appreciably affect
the calibration.
While the screen electrode provides a practically uniform electro-
static pressure over the surface of D, it is rather complicated to
compute the effective absolute values of h and of the electrostatic
area. This is more easily done by comparing it at low frequencies (say
at 100 c.p.s.) with a steel plate electrode in which the perforations
take up about 12 per cent of the total area. The surface facing D is
carefully machined so that h is uniform and known within less than
± 2 per cent. This is for absolute values of h in the range 0.075-0.080
cm. The acoustic load which this electrode imposes on D, with G
removed, is negligible at low frequencies. A lower limit on the
electrostatic correction for the perforations is made by adapting the
calculation given by Maxwell (" El. Mag.," 3d ed.) for rectangular
grooves in one plate of a parallel plate condenser. The above value of
R is corrected to
114 BELL SYSTEM TECHNICAL JOURNAL
1
R' = R-
J _ ^ . g ''^i gh
when
V5i' log, 2
6' A + g S (//+g)2
Si = area of perforation, 5 = total area.
II
An upper limit on the above correction is given by:
1
R' = R'
-f
The R' actually used was the mean of the above two values. The
value of R obtained with the screen electrode is shifted up or down to
make it coincide with R' given by the perforated electrode, at 100 c.p.s.
Appendix V
For frequencies below about 5,000 c.p.s. the difference between the
pressure and normal field calibrations is mainly due to two effects: (1)
reflection from the transmitter face and from the diaphragm; (2) air
resonance caused by the recess in front of the diaphragm.
Consider Fig. \A. Assume that in the circular aperture PQ, the
air particles are all moving in phase and parallel to AP. Then we may
treat PQ as a rigid massless piston in the wall RS. If RSjX is large
enough, the pressure on PQ held motionless will be double that of the
field pressure. The motional impedance of PQ imposed by the air
above P(3 is given by Rayleigh (Sound, vol. II, §302). Per unit area it is
where
Z] = pC{a + ib),
a = l-^^: b=-^K,(2kR): k = 'f: 2R = PQ.
Let Rp/Rp represent the ratio of " field " to " pressure " calibration.
Using the expression for plane wave propagation in a tube (e.g.
Crandall, Theory of Vibrating Systems and Sound, p. 99) we have at
once:
Rf _ . 1
„ — Z • ■-7-. ,
^ [cos kl + i(a + ib) â– sin kQ + ^Lia + ib) cos kl + i sin ^/]
where Za is the equivalent impedance per unit area of the transmitter
diaphragm, and / = AP. On substituting numerical values, Rp/Rp is
ABSOLUTE CALIBRATION OF CONDENSER TRANSMITTERS 115
found to have a maximum value of nearly Z.Z at w/2n = 3500 c.p.s.
This means that the air resonance adds a factor of 1.65 to the ordinary
doubling of pressure caused by a plane wall. A substantially similar
calculation has been given by W. West.^'
The observed RfIRp is a maximum at 3,500 c.p.s. but its value is
somewhat larger — nearly 3.65.
Bibliography
1. H. D. Arnold and I. B. Crandall, Phys. Rev., X, 22 (1917). E. C. Wente,
Physical Review, July 1917 and June 1922.
2. E. C.'Wente, Physical Review, April 1922.
3. W. West, Jl. I. E. E., Sept. 1929.
4. E. Gerlach, Wiss. Veroff. Siemens-Konz., Ill, 1923.
5. E. Meyer, Zs. F. Tech. Phys., p. 609, 1926; E. N. T., 4 1927; C. A. Hartmann,
E. N. T., 7, H. 3, 1930.
6. M. Grntzmacher u. E. Meyer, E. N. T., 4, 1927.
7. W. Zernow, Ann. d. Physik, 26, 1908.
8. Mallett and Dutton, Jl. I. E. E., May 1925.
9. L. J. Sivian, Phil. Mag., March 1928,
10. I. B. Crandall and D. Mackenzie, Phys. Rev., March 1922.
11. A. J. Aldridge, P. 0. E. E. JL, October 1928.
12. S. Ballantine, Phys. Rev., 32, 988, 1929; Proc. I. R. E., July 1930.
13. W. West, //. /. E. E., April 1930.
14. Rayleigh, Theory of Sound, Vol. II. ^ r^
15. L. J. Sivian, A Telephone Transmission Reference System, Elect. Comm., Oct.
1924. W. H. Master and C. H. G. Gray, Master Reference System for
Telephone Transmission, Bell Sys. Tech. Journ., July 1929.
Rating the Transmission Performance of Telephone
Circuits
By W. H. MARTIN
This paper discusses the rating of the transmission performance of
telephone circuits on the basis of the rate of repetitions in telephone conver-
sations and presents the rating method set up on this basis, which is being
adopted in the Bell System for determining and expressing the data for
the transmission design of the telephone plant.
A METHOD of rating the transmission performance of telephone
circuits is of course an essential in specifying the grades of trans-
mission service to be furnished, in designing, constructing and main-
taining telephone systems to provide the desired grades of service
economically and in the development of the various elements of the
telephone system which affect its transmission. As the art of tele-
phone transmission has developed and greater refinements have become
possible and desirable, changes have naturally been made in the
methods of specifying and rating transmission performance. Since
many such changes have been made in recent years, it seems opportune
to discuss the rating of transmission performance and to set forth the
rating method which is now being adopted in the Bell System for
determining and expressing the data for the transmission design of the
plant. In this connection, various methods which have been em-
ployed for measuring the transmission performance of telephone circuits
will be discussed to indicate their application and also their relation to
the new rating method. It is the purpose here to discuss this rating
matter primarily from the qualitative standpoint rather than to present
in quantitative detail the various relations involved in rating telephone
transmission. Obviously, the determination of many of these relations
presents sufficient material for separate treatment.
In carrying on a telephone conversation three major functionings are
involved, namely, that of the talker in formulating his ideas and utter-
ing words to convey these ideas, that of the telephone circuit in taking
the sounds of these words and reproducing them at another point, and,
lastly, that of the listener in hearing and recognizing these reproduced
sounds and in comprehending the ideas which they are intended to
convey. It is evident that all three of these functionings affect the
success of the telephone conversation. Since, however, the function-
ings of the talker and listener are common to both direct and telephone
conversations it might seem that the consideration of the transmission
116
RATING TRANSMISSION PERFORMANCE 117
performance of a telephone circuit could be limited to the functioning of
the circuit itself. In line with this, there has been some tendency to
confine such considerations to relations between the sounds reproduced
by the telephone circuit and the sounds impressed upon it. The per-
formances of the talker and listener, however, are materially affected in
certain important respects by the telephone circuit, and determinations
of the relative merits of the transmission performances of different
telephone circuits, must therefore go farther than the performances of
the circuits themselves and take account of the combined action of the
talker, circuit and listener.
Characteristics of Conversation
In view of this reaction of the telephone circuit on the talker and
listener, attention is directed to the pertinent characteristics of their
performances in both face-to-face and telephone conversations.
Direct Conversation
In direct or face-to-face conversations both the talker and listener
more or less subconsciously adjust their actions in many respects to
each other and to their circumstances. The loudness of talking is
placed initially at a level which experience has shown to be suitable for
the conditions and for the particular listener. If the listener indicates
verbally or by his expression that he is understanding easily or with
difficulty, some further adjustment may be made in the loudness of
talking. Since the talker judges his own talking level largely by the
loudness with which he hears his own voice, this level will be a function
of the amount of reverberation in the place in which he is talking.
Apparently for a wide variation of loudness in the customary talking
range, the speaker is not in general conscious of the amount of energy
which he is expending. Noise at the place of conversation also plays a
part in determining the talking levels since it makes louder talking
necessary in order to permit a given degree of understanding on the part
of the listener.
Along with an adjustment in talking level, the talker may improve
his enunciation if difficulty of understanding is expected or indicated
by the listener. There may also be a change in the manner of express-
ing the ideas in avoiding words which experience has shown are difficult
to understand and an idea may be stated in more than one way in order
to insure its comprehension. Other adjustments on the part of the
talker may be determined by his opinion of the mental acuity of the
listener, by the familiarity of the listener with the matter under dis-
cussion and by the interest in it. These factors affect the way of ex-
118 BELL SYSTEM TECHNICAL JOURNAL
pressing an idea, the kind and number of words used and hence the
time taken.
The Hstener also adjusts himself to the conditions by an amount
which is determined somewhat by his interest in the matter under
discussion. He may strive to comprehend the transmitted ideas and
require few repetitions by the speaker or he may refrain from exerting
himself and so tend to evoke greater effort on the part of the talker.
At times he may pretend not to understand in order to get confirmation
of a statement or to gain time in replying to a question.
In view of these factors and of the normal variations of different
talkers and listeners in all these respects, the portion of the questions
and statements of conversation which is correctly understood and the
time required to interchange certain ideas may vary widely for different
conversations even when they are carried on under a fixed set of local
conditions. If it were desired to determine a measure of the conversa-
tional satisfactoriness of these conditions, in addition to some quantita-
tive method for rating each conversation, there would be required,
therefore, observations on a large number of conversations between
different people in order to take account of the variables due to the
material of conversation, the people, and their abilities and desires to
accommodate themselves to the conditions.
Telephone Conversations
In telephone conversations, there are adjustments between talker
and listener as is the case in direct conversations, but there are certain
definite differences in this regard because of the interposition of the
telephone circuit between the participants. Here also, the speaker
tends to adjust his talking level to the loudness with which he hears his
own voice. In this case, however, he hears his own voice not only
through the air path, but also through the "sidetone" of the telephone
set, that is, through the electrical path from his own transmitter to his
own receiver. When this electrical path is more efficient than the
acoustic path, the sidetone will tend to control the talking level. It
has been found that varying the sidetone of the set has on the average
a definite effect on the talking volume of the speaker, the talking
volume being lowered as the efiiciency of the sidetone path becomes
greater.
In a telephone conversation, there is also a tendency for a person
in talking to adjust his volume on the basis of the loudness with which
he hears the person at the other end of the circuit. If the voice of the
other person comes through weakly, he judges that the connection re-
quires loud talking and acts accordingly. If the listener indicates that
RATING TRANSMISSION PERFORMANCE 119
he is not understanding, the talker may talk more loudly or closer to
his transmitter, and also make such adjustments in enunciation and in
setting forth his ideas as in the case of direct conversation. Also, the
loudness of talking may be affected by the room noise at the location
of the speaker, which noise incidentally may not reach the listener and
so play no part in his reception. Aside from cases where the room
noise at the far end is severe enough to be heard over the telephone
circuit, the speaker does not have definite knowledge of the room noise
at the listener's end and therefore is not in a position to adjust his
manner of talking to this condition except in so far as the listener may
indicate difficulty in understanding.
In listening, the result is of course dependent upon the position of
the receiver with respect to the ear. The local room noise reaches the
ear to which the receiver is held both by the path between the ear cap of
the receiver and the ear, and also through the sidetone path of his tele-
phone set. Some telephone users have learned that this effect may be
reduced by holding the receiver tightly to the ear and by covering the
mouthpiece of the transmitter when they are listening.
It is evident that the success of telephone conversations depends not
only upon the performance of the telephone but also upon the perform-
ances of its users, the material of their conversations, the way in which
they talk into the transmitters and hold the receivers to their ears and
the room noise conditions. In addition, it is seen that the performance
of the telephone affects the performances of the users in such important
respects as the loudness of talking, the manner of presenting the ideas
and the amount of effort exerted to understand. Also, the effect of the
room noise is a function of the circuit characteristics. Furthermore,
the reactions of the circuit performance on those of the users are not
constant but may vary from person to person and from conversation to
conversation. In view of the random nature of these factors, which
are beyond the control of those who design and operate the telephone
system, the service performance rating of a telephone circuit should be
on a basis which takes adequate account of their ranges and combina-
tions in practice. This points to a rating based on a statistical analysis
of results obtained under service conditions.
To determine and specify these factors so that it may be known how
to duplicate the range of service conditions in laboratory investigations
would be a prodigious task. Moreover, the duplication of these condi-
tions under control is bound to introduce a large element of artificiality
which would vitiate the results or at least raise serious questions as to
their dependability.
The practical solution is to get sufficient data regarding the results
obtained over telephone circuits of different performance characteristics
120 BELL SYSTEM TECHNICAL JOURNAL
by their normal users in carrying on regular conversations. This re-
quires a suitable quantitative method of rating conversations and
observations on a sufficient number of conversations over each circuit
condition to be investigated to constitute a reliable sample. This does
not mean necessarily that all the practicable circuit conditions have to
be observed in this manner but rather that sufficient data be so ob-
tained for the establishment of correlations with performance measure-
ments which are susceptible to laboratory determination. The funda-
mental point is that service performance ratings need to be based on
service results in order to take proper account of all the factors in-
volved.
Transmission Performance of Circuits
The distinction has been made between two kinds of transmission
performance of a telephone circuit, namely, that indicated by relations
between output and input sounds and that indicated by the results ob-
tained by the users of the telephone in carrying on their conversations
under service conditions. Performance indications of the first kind
will be referred to as "transmission characteristics" of the circuit. The
second kind of performance may be termed "transmission service per-
formance." The distinction between these two kinds of performance
is an important one and should be kept clearly in mind.
The output sounds dealt with in transmission characteristics are not
only the reproduced sounds which correspond to the input sounds but
also the accompanying extraneous sounds which are delivered by the
circuit. Also, the output sounds to be investigated cover not only
those delivered by the receiver at the far end of the circuit but also
those reproduced by the receiver in the station set containing the
transmitter energized by the input sounds. The sounds from the near
receiver include both those transmitted through the sidetone path of
the set and those returned to the sending end by reflection at impedance
irregularities in the circuit. Due to the time required for propagation
over the circuit these latter sounds may be delayed with respect to the
sidetone and hence appear as echoes. Likewise, echo sounds may be
delivered at the far receiver.
Transmission characteristics do not in themselves show the service
performance as realized by the users of the telephone but are essentially
indications of the functioning of the circuit in reproducing sounds.
They provide, therefore, a means for investigating and specifying the
performance of a telephone circuit without involving many, and in some
kinds of transmission characteristics any, of the actions of the talker
and the listener in conversation. With the establishment of proper
RATING TRANSMISSION PERFORMANCE 121
correlations between transmission service performance and transmis-
sion characteristics, these latter can of course be used to indicate
service performance.
In addition to specifying any kind or grade of circuit performance on
the basis of performance results there is the method, which has had
important practical application, of indicating performance in terms of
types of instruments and circuits and of the conditions of their use.
For example, a statement of the types of transmitters, receivers,
station sets and cord circuits and of the length and types of loops and
trunk, together with specific conditions of use, provides an indirect
specification of a performance. This method, which is extensively
used in many fields, may be termed the "instrumentality designation
method." An outstanding application of this method in telephone
transmission work is the Standard Cable Reference System ^ which
was so widely employed to provide a scale of performance. This
method has many present applications where physically determined
characteristics are unavailable or are difficult of definite determination
and specification. Also, the designation of instrumentalities is con-
venient in many cases because it provides a ready means of specifying
a practical combination of various kinds of transmission characteristics.
While this method is often expedient practically, taken by itself, it is
inherently cumbersome for the development of improved instrumental-
ities because of the lack of physical indication of the features to be
investigated.
Transmission Characteristics
The usefulness to the listener of the speech sounds reproduced over a
telephone circuit is a function of their loudness, of their distortion or
degree of departure from facsimile reproduction, and the magnitude
and character of the extraneous sounds or noise which accompany
them. Transmission characteristics are therefore directed primarily to
indications of the effects of the circuit and its parts on the reproduction
of sounds in these three respects. As already indicated, transmission
characteristics are determined not only for the path from transmitter
at one end to receiver at the other, but also for the sidetone and echo
paths.
Speech sound transmission characteristics, that is, expressions of the
relations between impressed and reproduced speech sounds, while they
have been extensively used, present some difficulty in quantitative
determination and specification because of their complex nature.
Also, the human element is involved in the persons used as generators
1 "Master Reference System for Telephone Transmission," Martin and Gray,
Bell System Technical Journal, July 1929.
122 BELL SYSTEM TECHNICAL JOURNAL
of the speech soundvS to be investigated and as observers to give indica-
tions of loudness and distortion and of their effects on the recognition of
the reproduced sounds. Two outstanding kinds of relations of this
type are those given by volume tests and articulation tests, which will
be discussed later. It has therefore been of great convenience to take
a further step and to study and specify the performance of telephone
circuits and their parts in terms of their functioning for single-frequency
sounds and currents. In this procedure, this functioning is investi-
gated for a number of different single-frequency sounds and currents,
so taken as to cover the range of frequencies transmitted by the circuit.
In the single-frequency transmission characteristics, the personal
element is eliminated and the measurements are made entirely on a
physical basis.
A great deal of attention has been given to the correlation of speech
sound and single-frequency transmission characteristics so as to enable
the former to be derived from the latter and so extend the application
of the type which is more readily susceptible to quantitative determina-
tion. Also, use has been made of easily specifiable multi-frequency
sounds and currents to permit the physical measurement to approach
more nearly speech sound conditions, of phonograph ic reproduction to
reduce the personal factor in the generation of speech sounds for meas-
urement purposes and of meter arrangements to simulate the ear rat-
ings of sounds, particularly from the standpoint of relative loudness.
As a result of the correlation of speech and single frequency charac-
teristics, extensive use has been made of determinations at selected
typical single frequencies to check the design, installation and main-
tenance of lines and other associated circuit elements.
The widely used volume test is essentially a means of specifying the
action of a telephone circuit or its parts, on the relation between the
reproduced and impressed sounds from the standpoint of their relative
loudness. In this test use has been made for many years of the
Standard Cable Reference System and recently of the Master Reference
System for Telephone Transmission ^ as references for comparison.
These reference circuits with their adjustable trunks provide a means
of obtaining different loudness ratios between input and output
sounds. By talking alternately over the reference circuit and the
one being investigated and adjusting the trunk of the reference
system until the output sounds of the two circuits are judged to be
equally loud, a specification of the loudness reproduction ratio is
obtained of the circuit under investigation in terms of the length of
the trunk in the reference system. The effect of a change in the
'^ See Reference (1).
RATING TR-iNSAIISSION PERFORMANCE 123
telephone circuit, such as the replacement of one receiver by another,
is measured in terms of the change required in the reference trunk
to give a loudness balance for the second condition. In this way,
measurements are also obtained of the effect on the loudness repro-
duction ratio of the various parts of telephone circuits. When the
circuits used commercially consisted of apparatus and lines similar to
those in the Standard Cable Reference System and the major con-
trollable factor was the loudness reproduction ratio, such measurements
constituted reasonably adequate means for indicating the comparative
functioning of circuits and apparatus.
The noise on a telephone circuit may be measured in various ways.
The method which has been most generally used is that of comparing
it with the controllable output of a fixed source of a complex wave shape
and adjusting this output until it and the line noise are judged to have
equal interfering effects.
With the availability of circuits and apparatus having widely differ-
ent distortion effects, the volume ratings became insufficient for indi-
cating the relative performances of commercial circuits. The earliest
method used in rating distortion effects was one in which observers
listening to transmission over the circuits, gave judgments as to their
relative merits. By so comparing various kinds and amounts of distor-
tion, two at a time, relative ratings can be established for placing them
in order of merit. This procedure was particularly useful in the early
days in working out the designs of transmitters and receivers, especially
from the standpoint of the location in the frequency range of their
points of maximum response. While such a judgment method has the
shortcoming of not providing quantitative ratings it has been found
that experienced observers can in general obtain results which are
relatively consistent with the results of more definite measuring
methods. Such judgment comparisons of distortion effects are fre-
quently used, particularly in exploratory work, and are still more or less
necessarily relied upon in setting limitations on circuit properties which
primarily affect the naturalness of reproduction.
To provide for the need of a method for measuring the relation be-
tween the reproduced and impressed sounds from the standpoint of
effects of different kinds of distortion, use has been made of the articula-
tion testing method.^' In this method, which has been widely used in
recent years, lists of syllables, usually meaningless monosyllables, are
called over the circuits to be rated and the percentage of syllables cor-
rectly understood is taken as a measure of the circuit performance.
3 "Articulation Testing Methods," Fletcher and Steinberg, Bell Sys. Tech. Jour.,
Oct., 1929.
124 BELL SYSTEM TECHNICAL JOURNAL
This testing method thus offers a means of indicating the distortion
effects of circuits in terms of the recognizabiHty of the reproduced
sounds of speech. Probably one of its first applications^ was in
determining the cutoff frequency to be used in the design of coil
loaded circuits.
The articulation testing method provides, of course, quantitative
measures in terms of the recognizabiHty of the reproduced sounds of
speech not only of distortion effects, but also of the effects of the loud-
ness of these sounds and of the noise which may accompany them.
This method has provided a very powerful tool for investigating the
effects of changes in the reproduction characteristics of telephone
circuits on the recognition of the reproduced sounds and has been par-
ticularly useful in indicating the lines to be followed in reducing causes
of distortion in circuits and apparatus and in evaluating the impair-
ment caused by noise on telephone circuits. It has been recognized,
however, that while such measurements indicate the capabilities of the
circuits in reproducing recognizable speech sounds, they do not in them-
selves give direct measures of the degree of success which the users of
the telephone obtain in conversations where their actions are free from
the control which is necessary in articulation testing and where the
contextural relation of the words plays such a large part in their recog-
nition. To make the results of this type of testing approach more
nearly the conversational results, words and sentences have been used
in place of the meaningless syllables but it is evident that even with
sentences, the control on the actions of the testers and on the ideas to be
communicated presents a condition which is quite different from those
of regular conversations.
All these ways of investigating and measuring the performance of
telephone circuits in reproducing sounds have useful applications in
present day transmission work. Frequently it is convenient to use
different methods for the various parts of a circuit in specifying the
complete functioning of the circuit in reproducing sounds.
Transmission Service Performance
From the standpoint of the users of the telephone circuit, the trans-
mission performance is measured by the success which they have in
carrying on conversations over the circuit. Different degrees of success
in this process may be taken as being indicated by the number of
failures to understand the ideas transmitted over the telephone and by
the amount of effort required on the part of the users to impart and
receive these ideas. Service performance is of course affected also by
* "Telephonic Intelligibility," Campbell, Phil. Mag., Jan., 1910.
IL4TING TILiNSMISSION PERFORMANCE 125
accidental irregularities in circuit conditions such as interruptions and
cutoffs, but from the standpoint of transmission design, attention can
be concentrated on the results obtained when the circuit is in normal
operating condition. Since failures to understand and exertion of
effort are experienced also in direct conversations, their occurrence in
telephone conversations obviously cannot be entirely ascribed to the
functioning of the circuit. Variations in these factors for different
types of circuits can, however, be used as a measure of the effect of the
differences in the transmission characteristics of these circuits.
The repetitions required in a conversation can be noted but a deter-
mination of the effort factor presents difficulties. There is undoubtedly
the tendency in carrying on conversations, as in other activities, to
exert no more effort than is necessary to obtain what the participants
consider to be satisfactory results. This effort, however, will in
general be increased as the difficulty of conversing becomes greater
and so bears a relation to the increase in repetitions. Also, it is prob-
able that two dissimilar circuits which cause the same rate of repeti-
tions when used for the same service, will, on the average, call for the
same amount of effort by their users.
In line with this, the rate of occurrence of repetitions requested by
the users of a particular telephone circuit in carrying on their regular
telephone conversations can be used as a direct measure of the service
performance of the circuit. By determining the repetition rate for a
large enough number of different people at the two ends to take account
of the variability of their personal characteristics in talking and listen-
ing to the telephone and of the conversational material and conditions,
a rating can be placed on the service afforded. By making such obser-
vations on connections having different transmission characteristics,
relative ratings can be established for these various transmission
characteristics.
It should be recognized, however, that while the rate of repetitions
required can be used for relative ratings of the transmission service
performance of different circuits, such ratings in themselves do not give
a complete picture of the service from the users' standpoint because
they do not show directly the amount of effort required. Some idea of
the effort exerted can be formed by the observers who are noting the
repetitions but this cannot be quantitative. In addition to the repeti-
tion rate and effort there is undoubtedly another factor which affects
the users' opinion of the service. In conversing over a circuit having a
poor transmission performance, annoyance or irritation may be felt by
the users because the amount of effort required may be considered by
them to be unreasonable. These factors, by their smallness or large-
126 BELL SYSTEM TECHNICAL JOURNAL
ness, may lead the users In the course of their conversations to make
favorable or adverse comments regarding the circuit performance.
These comments can be noted by the repetition observers and used,
together with any notations on effort and annoyance, to supplement the
repetition rating in arriving at a better picture of the service.
Effective Transmission Ratings for Plant Design
To provide for the transmission design of the telephone plant along
the lines of the previous discussion, ratings, termed "effective trans-
mission" ratings, are being determined which are based on the repeti-
tion rate in normal conversations. Circuits of different transmission
characteristics are considered to have the same effective transmission if
their repetition rates are equal when they are used for the same kind of
service. Furthermore, two changes in the transmission characteristics
of a circuit are taken as equivalent on the same basis. The effects of
such changes, however, are a function of the initial transmission char-
acteristics and it is therefore desirable to take as a basis for rating such
changes, a circuit which has characteristics typical in the various
respects of the ranges encountered in practice.
As a standard reference circuit for determining an expressing effective
transmission ratings, it is proposed to use a modification of the Master
Reference System, inserting in this certain amounts of distortion,
sidetone and noise to give it transmission characteristics comparable to
those of present commercial circuits. Pending the development of this
standard reference circuit, however, use will be made of a circuit con-
sisting of station sets and instruments of kinds in general use, loops of
typical length and construction connected by typical cord circuits to a
trunk of specified transmission characteristics. For this latter it is
convenient to assume a trunk having a cutoff typical of the loading
systems in use and having a frequency characteristic which is flat below
the cutoff point. It is also convenient to assume that the attenuation
of this trunk can be varied uniformly for all frequencies below the cutoff
point. This circuit may also be assumed to deliver at the two ends a
typical amount of line noise and to have typical room noise at the
terminals. Such a circuit then specifies a complete combination of
transmission characteristics which are typical of the telephone plant in
commercial use and may be considered as a working reference circuit.
The transmission service performance of such a circuit in commercial
use can be changed by varying the attenuation of the trunk and this
attenuation, expressed in decibels with respect to some reference value,
can thus be taken as constituting a scale for expressing different grades
of service performance.
RATING TRANSMISSION PERFORMANCE 127
Starting with such a circuit, changes can be made in its transmission
characteristics such as varying the attenuation of the trunk and its cut-
off, varying the length and type of the subscribers' loops, using different
types of transmitters and receivers in order to get different efficiencies
and kinds of distortion and changing the type of station circuit to get
different amounts of sidetone. By using circuits of these various char-
acteristics in commercial service and determining the repetition rates
obtained, a relation can be established between grade of service and
transmission characteristics both for different overall circuit combina-
tions and also for the various changes which can be made in such a
circuit. An outstanding advantage of selecting the type of circuit
which has been indicated, as a w^orking reference circuit, is that it
readily permits direct comparisons of the service performance of the
working reference circuit, or of circuits having closely similar charac-
teristics, with the service performances of various types of commercial
circuits.
It is desirable to go one step further and to express the effects of
changes in various transmission characteristics all in terms of changes
in some one characteristic of the circuit. For this latter has been
chosen the attenuation of the trunk. In accordance with this, then,
the effect of such things as differences or changes in cutoff of the trunk,
line noise, room noise, transmitter and receiver volume efficiencies and
distortions, sidetone, and, in fact, of any transmission characteristics of
any part of the circuit can each be expressed in terms of an equivalent
change in the attenuation of the trunk on the basis of equality of effect
on service performance. Thus the ratings of all such effects can be
placed on a basis which makes them readily comparable. For the
practical range of variations in these factors it has been found that in
general the effects so expressed can be added together with a good de-
gree of approximation. Where this is not the case, interrelated sets of
effective transmission ratings can be supplied to cover the various
typical combinations which are likely to be found in practice. This
places the application of the ratings given by this method on a com-
parable basis with the application of the old volume ratings, that is,
the assignment of a number to each part of the circuit, which numbers
can be combined by algebraic summation in arriving at an overall
rating for any particular circuit.
In line with this, the effective transmission of a trunk, for example,
is rated in terms of an attenuation loss of so many db plus a rating in
db which expresses the effect of the range of frequencies transmitted
with respect to some range selected as standard, plus another rating
expressed also in db to take account of the noise on this trunk. Simi-
128 BELL SYSTEM TECHNICAL JOURNAL
larly, loop loss curves can be drawn up for the combination of instru-
ments, set, loop and cord circuit such as has been used in the past on a
volume basis. On the new basis, these curves will include not only the
ratings of volume losses but also the ratings for the distortions in the
loop and instruments and the effect of the sidetone on transmitting and
receiving. In this manner, the transmission design of the plant can be
carried out in about the same manner as it has been on the volume rat-
ing basis but the effects of distortion, noise and sidetone can all be
included in these effective transmission ratings which are based directly
on service performance.
This in outline is the method of determining effective transmission
ratings which is now being worked to, its method of formulation and its
application. The complete discussion and description of these matters
involves innumerable details which, as already stated, it is not the pur-
pose to set forth here. From this outline it is seen that this method
provides the following outstanding things:
1. A scale for indicating different grades of effective transmission,
which scale is expressed in decibels and is directly correlated
with service performance by means of a typical circuit selected
as a reference. This permits the specification of grades of
service.
2. The use of this same scale as a means of assigning to each element
of practical telephone circuits an index, expressed in decibels,
which measures its contribution to the effective transmission of
the circuit, these indices being of such a nature that those cor-
responding to the elements in a circuit can be combined in a
simple way to give an overall performance index for that circuit.
Such a system of indices is necessary for plant design.
3. A means of correlating effective transmission service and circuit
transmission characteristics. This correlation is advantageous
in setting up the indices of (2) and in development and design
work in determining the desirability of possible changes in the
performance of the various elements.
The selection for the present of the typical practical circuit described
above, as a working reference circuit, has two important advantages,
which will be restated. First, by using a reference circuit having
typical transmission characteristics, the indices established for changes
in the various characteristics within the range of practical interest, are
directly applicable to the present plant and can be combined in a
simple manner to provide an overall circuit index. Second, and by no
means of minor importance in the earlier stages of the application of the
RATING TRANSMISSION PERFORMANCE 129
rating' method here described, by using as a reference a practical circuit,
it is possible and practicable to make direct comparisons of the service
performance of the reference circuit, or circuits having closely similar
characteristics, with the performances of various commercial circuits.
The maintenance of the first advantage will require, however,
changes in the working reference circuit as material improvements are
made in the transmission characteristics of the commercial circuits.
To obtain the second advantage means the use at present of carbon
transmitters in the working reference circuit. These are open to the
same objection here as they were in the Standard Cable Reference
System, namely, the difficulty of exactly specifying their performance
raises questions as to the reproducibility of their performance from
time to time. This was one of the major reasons for the replacement
of the Standard Cable Reference System as the basis for volume ratings
by the Master Reference System for Telephone Transmission with its
specifiable performance. To preserve the first advantage mentioned
and at the same time to obtain a reference system whose reproducibility
can be assured, it is the purpose, as more complete correlations are ob-
tained between transmission characteristics and service performance,
to associate with the Master Reference System, the means to make its
transmission characteristics meet the requirements necessary' to retain
the first advantage. Meanwhile the Master Reference System will
continue its function as a reference for volume ratings.
Determination of Ratings
To provide the basis for such a system of effective transmission
ratings as has been outlined, several series of tests hiive been made, the
most comprehensive of which has been under way for more than a year
between several hundred stations in the American Telephone and
Telegraph Company headquarters building and a similar number of
stations of the Bell Telephone Laboratories, between which there is a
large amount of intercommunication. The connections between these
stations are handled over special trunks in which the attenuation, cut-
off frequency and line noise can be varied. At the stations, different
types of instruments and station circuits have been employed. Ob-
servers are connected to each of these trunks who monitor the conversa-
tions over them and note the number of repetitions requested in each
conversation and also the duration of the conversation. In this way is
determined the repetition rate for a number of conversations between a
number of different people for the various combinations of circuit
characteristics so provided. Thus ratings are established directly of
such effects as those of trunk cutoff, noise on the trunk, different types
130
BELL SYSTEM TECHNICAL JOURNAL
of transmitters and receivers and of variation of sidetone in the station
set. In addition to the observations of repetitions, measurements are
made of the talking levels on the trunks by means of volume indicators
to determine the reaction of the circuit performance on talking levels.
An illustration of the results of such observation is given in Fig. 1.
The curve shows the variation of the repetition rate with change in
trunk attenuation for connections having the same kinds of terminal
sets and loops at both ends. This then provides a means of expressing
different grades of service performance in terms of trunk attenuation in
this circuit.
On this figure is shown also the repetition rate obtained for trunks of
two different effective upper cutoff frequencies. The change in trunk
4
/
3
y
2
C
>>
/
B©^
^
A = DISTORTIONLESS TRUNK
B=2700-CYCLE CUTOFF TRUNK
C = 1700-CYCLE CUTOFF TRUNK
-^
^
30
5 10 15 20 25
TRUNK EQUIVALENT AT 1000 CYCLES
Fig. 1— Relation between repetition rate and trunk equi\alent
attenuation required to produce the same increase in repetition rate as
that obtained in going to point C, for example, from the corresponding
1,000-cycle attenuation point on Curve A , is taken as the rating in db of
the lower cutoff' frequency represented by point C. This rating is
about 5 db. The rating of point B with respect to A is obtained in a
like manner to be about 1 db and correspondingly the rating of the cut-
off frequency of C with respect to B is about 4 db. This illustrates the
manner of obtaining effective transmission ratings for any change from
the characteristics of the circuit of Curve A.
It is obviously laborious to cover the ranges of all the transmission
characteristics of circuits of this kind. The idea is to establish points
which will cover the practical range and to use the results of articulation
RATIXC TRANSMISSION PERFORMANCE 131
tests and other similar measurements for interpolating between the
points established by the repetition method. In this way it is planned
to put the rating of transmission on a basis which indicates the effect on
service of changes in the various parts of the circuit.
Conclusion
In concluding, it may be restated that the primary purpose here has
been to discuss the rating of the transmission performance of telephone
circuits and the method which is being adopted in the Bell System for
determining and expressing effective transmission ratings for the design
of the plant. The salient features of this method which should be
emphasized are the following :
1. In establishing the rating of the transmission performance of a tele-
phone circuit, its performance is taken as that obtained when
the circuit forms part of the combination of talker, circuit and
listener, where the talker and listener represent the users of the
telephone system in commercial service.
2. The ratings of the effective transmission of circuits are based on the
rate of repetitions required.
3. The ratings of effective transmission will eventually be referred to a
modification of the Master Reference System arranged with
typical distortion, sidetone and noise. For a working reference
circuit, use is made of a circuit which has transmission charac-
teristics typical of those encountered in service. The trunk of
this circuit is taken as adjustable in attenuation for the purpose
of providing a scale for specifying different grades of overall
transmission performance and also for expressing ratings of the
effect on transmission performance of the various transmission
characteristics of circuits and their parts.
Paragutta, A New Insulating Material for Submarine
Cables *
By A. R. KEMP
Gutta percha and balata have proven eminently suitable for the in-
sulation of long deep sea telegraph cables, but their dielectric losses are too
high to meet the requirements of submarine telephone cables designed to
operate over long distances or of shorter cables employing carrier currents.
This paper describes a new material called paragutta which has been
developed to meet the present needs. It consists essentially of the purified
hydrocarbons of balata (or gutta percha) and of rubber together with minor
quantities of waxes to modify the mechanical characteristics. The puri-
fication of rubber particularly with respect to nitrogenous constituents
is necessary to effect electrical stability in water, A commercially usable
method of purifying rubber is described.
Evidence is furnished that paragutta has all of the desirable thermo-
plastic and mechanical properties of gutta percha while possessing such
superior insulation characteristics as to make it suitable for use on long
cables designed for transoceanic telephony. Its use is also advantageous
on shorter deep sea cables designed for carrier telephony as well as for
ocean telegraphs.
FORMERLY deep sea cables were used exclusively for telegraph
purposes but in recent years there has been an increasing use of
this type of cable for telephone service. Telephonic communication
requires cables of very much superior transmission quality to that
needed for telegraph. At the higher frequencies of voice transmission
the energy losses in the insulating material become a serious factor
and a radical improvement in submarine insulation is called for.
The longest existing deep sea cables operating at voice frequency
only slightly exceed 100 miles and the construction of a transoceanic
telephone cable with standard materials has been regarded as beyond
the practical limits of feasibility.
The installation and rapid expansion of transatlantic radio telephony
during the past few years have created a need for a deep sea telephone
cable to supplement this service, particularly during periods of at-
mospheric disturbances. In addition the development of carrier
telephony offers possibilities for increasing the trafific over shorter
submarine cables. For the shorter cable, the still higher frequencies
of carrier telephony make demands upon the insulating material
similar to those of long cables operating at voice frequency.
In view of these circumstances an extended study was undertaken
of the causes of losses and other electrical weaknesses of submarine
insulation and a search has been made for better materials. As a
* Jour. Franklin Instilule, Jan., 1931.
132
PARAGUTTA, A NEW INSULATING MATERIAL 133
result of this investigation an insulation called paragutta has been
developed which, as the name suggests, is derived essentially from
rubber and gutta percha. It is the purpose of this paper to describe
this material and give an account of the tests to which it has been sub-
jected to determine its suitability for the purpose.
By virtue of its superior electrical properties, the use of paragutta
in place of gutta percha for the insulation of telephone and telegraph
cables offers advantages either from the standpoint of improved trans-
mission or the economies in materials of construction which can be
made as a result of modified design.
Gutta percha and balata have been the standard materials for the
insulation of deep sea cables since the inception of the submarine
cable industry some seventy-five years ago. Although these sub-
stances are inadequate for modern telephone needs as regards their
electrical characteristics, their mechanical properties are peculiarly
adapted to submarine insulation. This is so much the case that gutta
percha can fairly be taken as a model which must be closely imitated
in respect to mechanical characteristics by any successful substitute.
This is fortunate since the use of any substitute which differs radically
from gutta percha would mean discarding large existing investments
in special technique, equipment and trained personnel, and would
involve serious risks as to the integrity of cables made with the new
material. It may be remarked in passing that no manufacturing
process requires a higher degree of insurance against occasional defects
than does the submarine insulation art, a fact that has engendered a
strong conservatism in the industry.
Because of its almost ideal mechanical properties, the requirements
for submarine cable insulation may conveniently be described by
reference to gutta percha. Gutta percha insulation, which often
includes more or less balata in its composition, is made of raw materials
carefully selected for quality, which are thoroughly washed and
extremely uniformly blended. The thermoplasticity of the material
is of great service in these operations and further permits it to be
readily extruded onto a conductor in multiple layers in a continuous
sheath with great exactness and freedom from mechanical defects.
After being forced around the conductor the material quickly sets to
a hard, tough covering when drawn through cold water. Its firmness
and toughness are essential to resist subsequent handling operations in
the factory, as well as those involved in laying, picking up and re-
pairing. The warm, soft material adheres readily to the conductor
and is well adapted to the making of joints in the insulation between
core lengths both in the factory and on the cable ship.
134 BELL SYSTEM TECHNICAL JOURNAL
In addition to those excellent and unique mechanical properties,
gutta percha possesses electrical characteristics peculiarly adapted to
submarine cable construction. Its outstanding electrical merit con-
sists in the fact that its electrical characteristics are stable under sea
bottom conditions over a great many years.
Gutta percha is obtained from the latex of a large number of species
of trees growing wild in the forests of the Malay Peninsula and the
East Indian Islands. The products of the various species of trees are
by no means of equal value, varying as they do in the content of
hydrocarbon, resins, moisture and other substances. Since the ma-
terial is gathered and worked up upon the spot by primitive people a
great deal of carelessness as well as deliberate adulteration is practiced
and the material comes upon the market in a dirty condition and in a
bewildering variety of forms which almost prohibit effective inspection,
standardization and grading.
The essential constituent of gutta percha is an unsaturated hydro-
carbon of colloidal nature which is similar in its chemistry to rubber.
It is this constituent which makes gutta percha plastic when warm
and tough when cold, and which contributes most conspicuously to
its electrical excellence as an insulator. The usual gutta percha in-
sulation is the result of blending and washing various grades of crude
gutta percha to remove dirt and water soluble components. The
hydrocarbon, resin, dirt and moisture contents as determined by
analysis of the crude material together with the electrical and mechan-
ical properties after washing are the principal characteristics used to
determine whether or not a particular grade of crude gutta percha is
suitable for use as submarine cable insulation. The hydrocarbon con-
tent of gutta percha insulation when applied to the conductor is usually
about 60 per cent, the remainder being mostly the natural resins to-
gether with small amounts of very finely divided dirt (humus) and
residual moisture. The proteins or albumens in crude gutta percha
and balata are almost completely removed by simple washing.
Balata comes from two species of trees of the same general botanical
family as gutta percha, but is native to the forest regions of upper
South America and is unknown in the gutta percha producing area
of the Far East. The latex of the balata tree is more fluid than that
of gutta percha, which permits the trees to be tapped and the fluid to
be collected at a central point in the forest, where the product from
various trees is mixed for recovery of the gum. Because of the small
number of species involved and the transportability of the fluid latex,
balata is produced in a much more limited number of grades and is
cleaner and more dependable as to uniformity of quality. Its essential
PARAGUTTA, A NEW INSULATING MATERIAL 135
constituent is the same hydrocarbon which gutta percha contains.
In addition to the hydrocarbon, there is present in balata some 40
per cent of resins and amounts of dirt, moisture and other impurities
which usually total about 15 per cent. The resins of balata are softer
than those of gutta percha and make the product in its raw state a
little less desirable than the better grades of gutta percha from the
mechanical standpoint. Balata, however, contains a smaller amount
of finely divided dirt or humus than gutta percha, which is reflected
in its superior electrical characteristics and lower water absorption.
The resins of both gums have been usually included with the hydro-
carbon in making submarine insulation. Sometimes, however, a
portion of the resins are removed, partly to increase the toughness and
partly to improve the electrical characteristics.
There are several methods which may be used for preparing gutta
hydrocarbon nearly free from resinous substances. One of these
methods involves dissolving the balata or gutta percha in warm
petroleum naphtha, filtering the solution from dirt and precipitating
the gutta hydrocarbon from solution by refrigeration, leaving most of
the resins in solution. A simpler and less expensive method, however,
is that of leaching out the resins by simply soaking the sheeted or
finely cut material in a suitable grade of petroleum naphtha at ordinary
temperature, followed by draining off the solution of resins and finally
evaporating the residual solvent from the extracted material.
The completely deresinated hydrocarbon from either source is not
suitable for use alone as submarine cable insulation because insufii-
ciently plastic at safe working temperatures, as well as prohibitively
expensive. Otherwise the complete deresination of these products
would be highly advantageous as, for example, is indicated by the
superior electrical characteristics of deresinated balata shown in
Table I. A substantial amount of experimentation upon the methods
of refining balata has been necessary to secure the excellent electrical
characteristics therein indicated but no revolutionary innovation has
been necessary.
TABLE I
Effect of Resin Content on the Electrical Characteristics of Balata
Electrical Characteristics 0° C, 1 Atm.,
2000 Cycles
Dielectric Specific Conductance
Material Constant Unit = 10"'= mho. cm.
Balata 3.1 66
Deresinated Balata 2.6 3
Balata Resins 2i.2> 52
In attempting to develop a new insulating material for deep sea
cables it seemed best to begin with gutta hydrocarbon as a basis,
136 BELL SYSTEM TECHNICAL JOURNAL
since its mechanical properties are so unique, rather than to attempt
to synthesize a new chemical compound which would imitate it. In
order to overcome the excessive stiffness of the pure gutta hydrocarbon,
as well as its prohibitive cost, it was determined to attempt to blend
large quantities of rubber with it, since rubber is the nearest kindred
material and is commercially available at low cost. There resulted
thermoplastic products of fairly good mechanical characteristics which,
however, proved to be insufficiently stable electrically.
Meanwhile, a thorough study was being made of the electrical and
physical characteristics of rubber and particularly of the causes of its
electrical instability upon prolonged immersion in water. Our hope
that such a study would not only reveal the nature of the defects of
rubber but also suggest means for remedying them has been realized
to a gratifying degree.
Rubber, as is well known, is also derived from the latex of certain
trees, chiefly Hevea Brasiliensis. This tree has been cultivated in
large areas on the plantations in the Far East and the product is
obtainable commercially in excellently standardized grades. Its
principal constituent is a hydrocarbon scarcely distinguishable from
that of gutta percha by chemical means, but radically different from
it in physical properties, notably in that it has but a slight degree of
thermoplasticity and is far more distensible in the cold state. Aside
from the hydrocarbon, rubber also contains small amounts of resins,
proteins and other impurities, but the aggregate non-hydro-carbon
constituents in the better grades are usually less than 10 per cent in
contrast to 50 per cent or thereabouts for gutta percha and balata.
Rubber is used almost exclusively in industry in a vulcanized form,
that is, in combination with a small percentage of sulphur. In this
form rubber has also been used to a limited extent for submarine cable
insulation, but has long been recognized as lacking sufficient electrical
stability for deep sea cables designed to carry a heavy traffic. It is
still used to a considerable extent with a fair degree of success for
insulation on short cables where the electrical requirements are not
severe. In tropical waters it has the advantage over gutta percha of
greater resistance to teredo attack and to damage by high temperature.
Some years ago an extended study ^ was made of the causes of the
electrical instability of vulcanized rubber, which led to the conclusion
that the water soluble impurities are largely responsible. These
impurities can be removed comparatively readily and satisfactorily
in the process of manufacture, and a submarine insulation of a fair
degree of stability is thereby attained.
^ Williams and Kemp, Jour. Franklin Inst., 230, 35 (1927).
PARAGUTTA, A NEW INSULATING MATERIAL 137
Even so, vulcanized rubber is very inferior to gutta percha for
submarine insulation as the necessary manufacturing operations are
more difficult and likely to lead to defects. The removal of mechanical
impurities is by no means simple because the raw stock is not plastic
enough for thorough straining. The lack of plasticity also interferes
with multiple covering of conductors, and the process of heating to
bring about vulcanization is liable to result in deformation of the
insulating layers. The joining and repairing of core lengths insulated
with rubber is also more of a problem than with gutta percha, which
can be so readily remolded in case imperfections appear in the course
of the process.
The methods of electrical stabilization of vulcanizable rubber
compositions are only partially effective in the absence of vulcanization
and it was therefore necessary to extend the study in an effort to
secure the desired electrical properties in rubber in the raw state.
It might be supposed that mere admixture of raw rubber with gutta
hydrocarbon would produce the necessary stability. This is true
only to a limited extent. When the proportions of rubber are high
enough to meet the mechanical and economic requirements, the
electrical stability is impaired.
Effect of Proteins on Electrical Stability of Crude
Rubber Immersed in Water
It has been previously shown that crude rubber contains consider-
able water soluble impurities and that their removal results in a
large reduction in water absorption. i- ^ Rubber so prepared absorbs
no more water than good cable gutta percha but in a raw state when
immersed in water, it fails sooner or later as an insulator, often sud-
denly and completely.
To determine the reason for this electrical instability of crude rubber
in water, samples of very pure rubber hydrocarbon completely freed
from proteins, resins and other impurities were prepared and tested.
It was found that this material not only absorbed very little water
but showed practically no change in electrical characteristics as a
result of prolonged immersion in water. The impurities natural to
rubber therefore seem to be responsible for its instability.
It has been known for many years that crude rubber contains
proteins, ordinary plantation rubber containing about 3 per cent.
Previous investigators have postulated and shown considerable
indirect evidence to the effect that the rubber globules in rubber latex
have an adsorbed film of protein around them and that this condition
2 Lowry and Kohman, Jour. Phys. CJtem., 31, 23 (1927).
138 BELL SYSTEM TECHNICAL JOURNAL
also exists in crude rubber. It is also known that latex serum contains
a substantial quantity of protein in solution. The preparation of
crude rubber from latex by addition of acid or by processes of evap-
oration of the water by heat undoubtedly results in the precipitation
of considerable quantities of this protein which becomes entrapped
between the globules as they coalesce. It is easy then to visualize
that in crude rubber there exists a continuous phase of protein or a
protein network which, acting like most protein matter, absorbs large
quantities of water, resulting in paths through which electrical con-
duction occurs.
Removal of Nitrogen Containing Bodies from Rubber
The problem of developing a suitable commercial method for
preparing rubber free from nitrogenous matter offered many apparent
difficulties. The proteins are colloidal in nature and in the presence
of water form gelatinous masses rather than true solutions. On this
account they often cannot be removed by simple washing as can be done
in the case of gutta percha and balata. It has been known for some
time that proteins can be broken down to water soluble products by
boiling with dilute hydrochloric or sulphuric acids. This treatment
did not produce satisfactory results in the case of rubber. As a result
of many experiments involving a variety of methods, it was found that
heating rubber in an autoclave at an elevated temperature in the
presence of water alone fairly rapidly brought about the desired hy-
drolysis of the rubber proteins, converting them to water soluble
materials. As a result of subsequent washing, it was found that the
nitrogenous bodies had been almost completely eliminated without
deleterious effect on the rubber hydrocarbon.
Rubber either in the form of sheets immersed in water or as an aque-
ous rubber dispersion such as latex can be employed in the process.
The treatment of latex, however, results in a more rapid hydrolysis
of the proteins. Considerable latitude exists in the choice of condi-
tions, but the following example will suffice to describe one method of
carrying out the process: ammonia preserved latex is diluted 1 to 5
with pure water. The latex is then heated in an autoclave for approx-
imately ten hours at 150° C. After cooling it is coagulated with
acetic acid and thoroughly washed. As a result of this treatment
the nitrogen content of the rubber is found to be less than 0.10
per cent, which is about one fourth that of ordinary plantation crude
rubber. Figures 1, 2, 3 and 4 illustrate the relative water absorption
and electrical stability of deproteinized rubber as compared with the
ordinary crude product. Vulcanized deproteinized rubber was also
PARAGUTTA, A NEW INSULATING MATERIAL
139
found to be somewhat superior to ordinary vulcanized crude rubber
as regards its electrical stability in water.
In addition to the superior electrical stability of deproteinized rubber,
it was found to be more readily plasticized and mixed with gutta
H 1.5
5 1.0
SMOKED
SHEET
^
/
)
/
SMOKED
SHEET-
/
/
/
^
'
DEPROTEINIZED
A
^
2 3
TIME IN WEEKS
Fig. 1 — Effect of washing and removal of protein on the water absorption of crude
rubber when immersed in 3.5 per cent NaCl solution at room temperature.
(? 35
t 25
1
\
1
1
>
;
1
1
/
/
/
/
y
/SMOKED
SHEET
,;
/ SMOKED
/ SHEET-
1 WASHED
/
/
/
/
l_
\
/
RUBBER
1 1
2 3
TIME IN WEEKS
Fig. 2 — Effect of washing and removal of protein on the dielectric constant of crude
rubber when immersed in 3.5 per cent NaCl solution at room temperature.
than is the case with crude rubber, thereby yielding a product with
better thermoplastic properties.
140
BELL SYSTEM TECHNICAL JOURNAL
Preparation of Paragutta
As previously stated, the principal constituents of paragutta are
deproteinized rubber and purified gutta hydrocarbon. Specially
treated hydrocarbon or montan waxes may also be added as a third
constituent to modify mechanical properties and reduce cost. The
proportions of these constituents may be varied over a wide range to
achieve the desired characteristics, but in general rubber and gutta
10'°
K
inl6
i
I \
\
DEPROTEINIZED
a.
\
\
s
^ RUBBER
i 10'^
y-
z
\
\
o
a.
\
\
^ 10"^
in
2
\
\ SMOKED
\ SHEET -
\ WASH ED
I
o
? lO'O
>
\
i
\
\
>
1-
iri
u
a.
\ SMOKED
\SHEET
\
\
\
,o6
\
\
\
1
\
1
10^
2 3
TIME IN WEEKS
Fig. 3 — Effect of washing and removal of proteins on resistivity of crude rubber,
when immersed in 3.5 per cent NaCl solution at room temperature.
are used in about equal proportions and purified montan wax may be
added up to about 40 per cent. Superior electrical properties, how-
ever, result from the use of hydrocarbon waxes, which may be added
in amounts up to about 20 per cent. By the proper blending of these
materials, a thermoplastic insulation is obtained which closely ap-
proximates gutta percha in mechanical properties and is fully its equal
as to electrical stability in water. Its specific electrical characteristics
PARAGUTTA, A NEW INSULATING MATERIAL
141
represent a substantial improvement over those of the classical insula-
tion compounds and its cost is lower.
The final steps in processing paragutta are very similar to those
used for gutta percha and involve blending and washing the depro-
teinized rubber and deresinated balata or gutta together, masticating
to remove excessive water and at the same time incorporating such
waxes as are found necessary. The material is then strained through
fine sieves under hydraulic pressure to remove adventitious impurities,
kneaded to remove air and finally placed on the covering machine rolls
to be forced around the conductor. The machinery in use for pro-
I 140
5
- 120
^
o
8 100
O 40
1
1
1
1
1
1
1
IsMOKED /
/sheet /
/ SMOKED
/ SHEET-
/ WASHED
/
I
DEPf
^OTEIN
IZED
u
/^
F
-1
UBBER
1 "
2 3
TIME IN WEEKS
Fig. 4 — Effect of washing and removal of proteins on conductivity of crude rubber
when immersed in 3.5 per cent NaCl solution at room temperature.
cessing gutta percha is suitable for handling paragutta in these
operations.
Comparative Properties of Paragutta and Gutta Percha
Tensile Properties: Although submarine insulation is not subjected
to tensile deformation in practice, tensile properties indicate to some
degree the relative mechanical suitability of a given material for the
purpose. Figure 5 shows the stress-strain characteristics of paragutta
and gutta percha submarine cable insulation. These results show
142
BELL SYSTEM TECHNICAL JOURNAL
that paragutta has tensile properties equal to cable gutta percha
although its gutta content is substantially lower.
Compression Properties: The insulated submarine cable conductor
commonly known as the core is frequently subjected to uneven com-
pression stresses during manufacture, laying and repairing. The
insulation must therefore be capable of withstanding these stresses
without appreciable deformation. To determine the relative merits
of paragutta and gutta percha in this respect their comparative stress-
strain characteristics under compression have been measured, using a
special compression machine,^ and are shown in Fig. 6. In this test
125
200 300 400 500
ELONGATION -PER CENT
Fig. 5 — Comparative tensile properties of paragutta and gutta percha at 25° C.
a steel rod 1.6 cm. in diameter was forced endwise into a sheet of the
material .375 cm. in thickness at a rate of about 4 cm. per minute
while simultaneously recording the deformation and load. These
results show that very little difference exists between these materials
in this test, and factory handling of cores confirms the general con-
clusion.
Flexibility: The flexibility of submarine cable insulation is important
because the core is subjected to considerable flexing during manu-
3 Hippensteel, Bell Laboratories Record, S, 153 (1928).
PARAGUTTA, A NEW INSULATING MATERIAL
143
facture, laying and repairing and possibly at times during use, espe-
cially where tidal currents may cause movement in the cable. Para-
gutta and gutta percha cores have been subjected to slow and con-
tinuous flexing at 0° and 25° C. for long periods and it was found that
both materials will withstand millions of repeated flexures at small
amplitudes without failure. When the amplitude of flexure was
increased to strain the conductor slightly beyond its elastic limit,
the conductor always failed in advance of the insulation.
Plasticity Tests: Laboratory tests were made to determine the rela-
tive plasticity of paragutta and gutta percha, using both the Williams ^
9 100
40 60
COMPRESSION-PER CENT
100
Fig. 6 — Comparative compression properties of paragutta and gutta percha at 25° C.
and the Marzetti ^ type of plastometers. These tests are valuable
guides but the final judgment of a material as regards thermoplasticity
was made by determining its workability on commercial gutta percha
insulating machines. Paragutta is somewhat more resistant to flow
than gutta percha at temperatures ranging from about 40° to 70° C.
When applied to the conductor, however, its greater resistance to flow
at elevated temperatures can be taken as an advantage as it lessens
the danger of faults occurring if the core should be accidentally exposed
to elevated temperatures or to conditions which might exist in con-
nection with cable used in the tropics.
* Williams, Jour. Ind. & Engg. Chem., 16, 262 (1924).
^ Marzetti, Giorn. Chitn. Ind. Applicata, 5, 342 (1923).
144
BELL SYSTEM TECLINICAL JOURNAL
Figure 7 shows the relative plasticities of cable gutta percha and
paragutta at several temperatures as determined by the Williams *
method, which can be taken to indicate the relative plasticities of
these materials at working temperatures.
Brittle Temperature: It is extremely important that the temperature
at which submarine cable insulation becomes brittle should be far
below the range of sea bottom temperatures to be encountered in use.
This is one of the properties in which rubber and gutta percha greatly
excel any other available insulating material. Kohman and Peek ^
1
1
\^
\
1 -CABLE GUTTA PERCHA
2 -PARAGUTTA
\
\z
\
\
V
\
\
\
s
\,
"^
\
X,
^»^
1
2
40
50
90
100
60 70 80
TEMPERATURE -DEG. C
Fig. 7 — Effect of temperature on the plasticity of cable gutta percha and paragutta.
have described an apparatus for accurately determining this tem-
perature. The brittle temperature of paragutta is somewhat lower
than cable gutta percha, as can be seen from the results in Table II,
which give the range of brittle temperature values found for different
samples of several materials.
Water Absorption — Electrical Stability
The amount of water absorbed by rubber and gutta percha when
immersed in water is the result of a complicated mechanism. The
quantity and nature of water soluble or water absorbing impurities
* Kohman and Peek, Jour. Ind. &" Engg. Chem., 20, 8 (1928).
PARAGUTTA, A NEW INSULATING MATERIAL 145
TABLE II
Brittle Temper.vture of Paragutta and Other Insulating AIaterlvls
Brittle Temperature
Material ° C
Gutta Percha (Cable Insulation) —23 to —36
Paragutta -45 to —61
Balata (Washed) -44 to -52
Balata (Washed and Deresinated) —62 to —67
Crude Rubber -57 to -58
Vulcanized Rubber (Soft) -53 to -58
in the rubber or gutta percha and the salt concentration of the water
in which the samples are immersed are controlling factors. The
enormous increase in the quantity of water absorbed by ordinary
rubber when immersed in distilled water as compared with its absorp-
tion in salt solutions has been explained on the basis of osmotic
theory.^ In accordance with this theory rubber acts as a semi-
permeable membrane. Water soluble crystalloids or hydrophillic
colloids (proteins) attract the water which enters the rubber by
diffusion. When immersed in distilled water these impurities tend
to reach infinite dilution with water, being opposed in this by the
resistance of the rubber itself to swelling. In salt solutions the
amount of water absorbed is finite and depends on the equalization
of osmotic pressures of the internal and external solutions. The
change in water absorption of pure rubber hydrocarbon with the salt
concentration of the external solution is small over the whole range,
which indicates that the water enters by a process of solution. This
has also been found to be the case for gutta hydrocarbon and is more
or less true for paragutta and gutta percha. The water absorption
in distilled water can therefore be taken as a measure of the freedom
from water soluble or water absorbing impurities. Figure 8 shows
the effect of NaCl concentration in the immersion solution on the
quantity of water absorbed by samples of rubber, paragutta and gutta
percha at room temperature. Samples of rubber containing water
soluble matter or proteins do not readily reach an equilibrium water
content in distilled water. Crude rubber has been found to absorb
more than 100 per cent water in distilled water at ordinary temperature
without reaching equilibrium. ^ Gutta percha, paragutta and pure
rubber hydrocarbon on the other hand reach a definite and lower
equilibrium water content in distilled water, which shows their greater
freedom from water soluble or water absorbing matter.
As the electrical stability of paragutta in sea water is of paramount
importance an exhaustive study has been made on a large number of
specimens as regards their changes in electrical values over long periods
of immersion in 3.5 per cent salt solution. Gutta percha insulation
146
BELL SYSTEM TECHNICAL JOURNAL
contains about one per cent water when at equilibrium with sea water
whereas paragutta contains somewhat less than this amount. These
values have been determined by testing samples made up with various
water contents below and above equilibrium values and determining
the water content after prolonged immersion in 3.5 per cent NaCl
solution, as seen in Figure 9. The equilibrium value is practically
the same when equilibrium is approached from either direction.
7
\
•-PLANTATION SMOKED SHEET RUBBER
X- WASHED SMOKED SHEET RUBBER
o-GUTTA PERCHA
A-PARAGUTTA
+-PURE RUBBER HYDROCARBON
1-
Z
\
(J
cr
UJ 4.
1
Z
1-
Q.
a.
<
'
\
^
*.
\
V
>
\^
\
^^
\^
^*^«->
s
^^
r^
'
^^^
H
^
^4:^
^
f
â– -^
=^
Fig. 8-
5 10 15 20 25 30 35
CONCENTRATION NaCl IN IMMERSION SOLUTION-PER CENT
-Relation of water absorption to salt concentration in immersion solution.
The overall quantity of water absorbed, however, cannot be used
as a final criterion by which to judge insulation for it has been pre-
viously shown (Figs. 1 to 4) that washed crude rubber completely
fails as an insulator after absorbing less than one per cent water. The
mode of distribution of water absorbing impurities in an insulating
material has been found to be of utmost importance as regards the
magnitude of the effect of moisture in various insulating materials.
Examples where large effects on insulating properties are caused as a
result of moisture absorption by localized impurities are found in the
above case of proteins in crude rubber, water soluble salts associated
PARAGUTTA, A NEW INSULATING MATERIAL
\\1
with tillers in vulcanized rubber ^ and hygroscopic salts on the surfaces
of textile fibers.^
On the other hand, the electrical properties of paragutta or gutta
percha are not impaired when several times their equilibrium water
content is incorporated with them. Gutta percha, however, does
show an increase in capacitance of about 10 per cent as a result of
water absorbed by a completely dried specimen, but as it is always
the practice to apply it to the conductor in a wet condition this
6 8
TIME-MONTHS
Fig. 9— Changes in water content of 50 mil wet and dry paragutta and gutta perclia
sheets when immersed in 3.5 per cent NaCl at room temperature.
change is not of practical significance. The electrical properties of
paragutta on the other hand show practically no changes as a result
of moisture absorption by a dry sample. These facts are taken to be
the best evidence of the electrical stability of paragutta in contact
with water.
Hundreds of specimens of paragutta and gutta percha have been
studied as regards changes taking place in electrical characteristics
after long periods of continuous immersion in 3.5 per cent salt solution.
These tests, some of which have been for periods of three to five years,
show that paragutta is fully equal to gutta percha as regards its
^ Williams and Murphy, Bell Sys. Tech. Jour., 8, 225 (1929;.
148 BELL SYSTEM TECHNICAL JOURNAL
stability. Wlien properly prepared both of these materials show-
practically negligible changes in electrical properties as a result of
prolonged submergence in water. Sea bottom conditions are even
less likely to affect these materials than those existing in the laboratory.
This is because of the absence of light, limited oxygen supply and
low temperature, all of which reduce the tendency of materials such as
paragutta or gutta percha to oxidize or otherwise deteriorate. It
has also been shown ^ that the low temperature and high pressure
existing at sea bottom reduce the rate of water absorption but do not
materially affect the amount absorbed.
Electrical Characteristics: The electrical properties of paragutta
depend upon the particular composition chosen, the quality of the
raw materials and the care exercised in processing them. For long
telephone cable insulation, it is necessary to exercise the utmost care
to obtain a material having dielectric constant and specific conductance
values sufficiently low to reduce to the minimum its effect on the
attenuation. On the other hand, for ordinary telegraph cables these
values are less critical and it may be advantageous to modify the
practice for purposes of economy. Representative values for the
electrical properties of a superior grade of paragutta and typical cable
gutta percha under sea bottom conditions are given in Table III. It
will be seen in this table that paragutta has a 20 per cent low^er dielec-
tric constant and a specific conductance one-thirtieth that of ordinary
cable gutta percha under sea bottom conditions. The insulation
resistance and dielectric strength of the two materials are practically
the same.
TABLE III
Comparative Electrical Properties of Paragutta and Cable Gutta Percha
AT Sea Bottom Conditions
Specific Inductive Effective A-C
Capacity 2° C. Conductivity 2° C,
400 Atm., 2000 Cycles 400 Atm., 2000 Cycles
Unit = 10-12 mho. cm.
Cable Gutta Percha 3.3 90
Paragutta 2.6 3
Acknowledgment
The author wishes to acknowledge his indebtedness to Mr. R. R.
Williams for counsel and assistance during the prosecution of the
work and writing of the paper.
Abstracts of Technical Articles From Bell System Sources.
An Efficient Loud Speaker at the Higher Audible Frequencies} L. G.
BosTWiCK. This paper describes a loud speaker designed for use as an
adjunct to existing types for the purpose of extending the range of
efficient performance to 11,000 or 12,000 cycles. A moving coil piston
diaphragm structure is used in conjunction with a 2000-cycle cutoff
exponential horn having a mouth diameter of about 2 inches. Mo-
tional impedance measurements on this loud speaker indicate an aver-
age absolute efficiency of about 20 per cent within the frequency range
from 3000 to 11,000 cycles. The variation in response within this
band does not exceed 5 db. By using a high-frequency loud speaker of
this type the efficiency and power capacity of the associated low-fre-
quency loud speaker can be improved and a uniform response-fre-
quency curve from 50 to 12,000 cycles can be obtained.
Results of Noise Surveys. Part I. Noise Out-of-Doors? Rogers
H. Galt. The purpose of a noise survey of a locality is to study the
space and time distribution of noise intensity, the frequency composi-
tion of the noise, the contributions of various noise sources, the relation
between the annoyance effect of the noise and its physical and auditory
characteristics, and the effectiveness of methods of noise reduction.
The extent to which each of these phases of the noise problem has been
investigated heretofore has depended upon the point of view of the
investigator and upon the apparatus employed. From one standpoint
or another, any audible sound may fall within the category of noise;
hence the variety of possible noise surveys is almost unlimited. Not
many such surveys have been carried out, however, partly because the
appropriate apparatus is of recent development ; nor has any extensive
comparison been published between the results obtained in different
places and with different instruments. It has therefore seemed worth
while to assemble such previously published results as are available,
and certain new observations, in the present series of papers, of which
this paper deals with noise out-of-doors.
Microphonic Action in Telephone Transmitters.^ F. S. Goucher.
This semi-technical article gives a brief resume of the theories of micro-
phonic action and describes the results of some experiments on the
1 Jour. Acous. Soc. Amer., July, 1930.
^ Jour. Acous. Soc. Amer., July, 1930.
3 Science, Nov. 7, 1930.
149
150 BELL SYSTEM TECHNICAL JOURNAL
contact behavior of granular carbon of the type used in commercial
microphones.
A technique is described whereby contacts — either singly or in
groups — may be studied under contact forces of the order of 1 dyne.
Through a study of the temperature coefficients of resistance of such
contacts it is possible to conclude that the conducting portions of the
contact junctions are of the nature of carbon and that new contact
points are established or broken when the resistance is varied in a
reversible resistance force cycle.
The experiments show that for such reversible cycles the relation
between the resistance and force is of the approximate form
R = K(F)~"'. The exponent n varies considerably from cycle to cycle
but its average value depends on the force limits. The largest values
of w are obtained with the aggregates of granules under such conditions
of force limits that the elastic strains must be relatively large. A
maximum mean value substantially independent of the force limits
over a wide range closely approximates the value 7/9.
This value 7/9 is the maximum given by a theory of contact resistance
worked out by F. Gray, assuming that the contact is made between
two spheres of conducting material having surface roughness equivalent
to an assembly of minute spherical hills. On account of the elasticity
of the material both the microscopic area of contact between the spheres
and the microscopic areas of contact between the hills increase with
contact force. A strained aggregate of granules may therefore be made
to behave like an ideal single contact between spheres having a rough
surface.
For single contacts and for aggregates at small strains the value of n
falls below the minimum value 1/3 which is accounted for by theory.
This is associated with internal contact forces, or cohesion, which render
the contacts relatively insensitive to changes in the applied force.
The existence of cohesion is readily demonstrated by the fact that
contacts always require a finite force to break them even when no
current has passed through the contact.
The Architecture of Living Cells — Recent Advances in Methods of
Biological Research — Optical Sectioning with the Ultra- Violet Micro-
scope} F. F. Lucas. In previous papers of the past few years the
development and application of the ultra-violet microscope to the
science of metallography have been described.
Metallography, at first thought, appears wholly unrelated to his-
tology or other branches of biology but the two branches of science do
* Proc. Nat. Acad, of Sciences, Sept., 1930.
ABSTRACTS OF TECHNICAL ARTICLES 151
have many points in common. Both deal in the last analysis with the
structure of matter and, in each, the microscope is an indispensable
tool. Improvements in microscopic vision which enlarge the world of
vision in one branch of science inevitably have a reflection in the other.
It is not the purpose of this paper to enter into a discussion of struc-
tures of living cells as revealed by the ultra-violet microscope. More
particularly, the object is to present a tool for biological research; a
tool which enables us to photograph the structure at different planes or
levels within a single cell or group of cells ; one which enables us to see
the living cell with a degree of precision and clarity not heretofore
possible by any other known means and with a potential resolving
ability at least twice that of the best apochromatic system using visible
light.
Production of Plastic Molded Telephone Parts. ^ A. M. Lynn. The
Western Electric Company now manufactures for Bell System ap-
paratus a large number of different phenol-plastic, shellac, and hard-
rubber molded parts, the output of which varies from a few thousand
to several million per year. The majority of these molded parts are
produced in comparatively small quantities, but certain of them, such
as the phenol-plastic molded parts used in the hand-set type of tele-
phone, a new molded subscriber's set housing, and the receiver shell,
cap, and mouthpiece used on the older type of desk-stand telephone,
are hea\y-running parts. The tools and press equipment used in the
production of these parts are described in this paper.
Variation of the Inductance of Coils Due to the Magnetic Shielding
Effect of Eddy Currents in the Cores.^ K. L. Scott. An analysis is
made of the shielding effect of eddy currents on the flux in the interior
of cores of cylindrical or flat sheet material. It is shown that the
counter voltage of self inductance of an iron-cored coil is due only to
the component of flux in the core which is in phase with the flux at
the surface of the core. Expressions are obtained and curves plotted
showing the variations of inductance of a coil with frequency, or with
the conductivity and permeability of the core material. Sample
calculations and some experimental results are given. The results
show that the inductances at high frequencies are actually less than
the predicted values, which leads to the suspicion that some factor
other than eddy currents causes the flux in the interior of the cores to
decrease with increasing frequency.
5 Mech. Engg., Oct., 1930.
6 Proc. I. R. E., Oct.. 1930.
152 BELL SYSTEM TECHNICAL JOURNAL
Results of Noise Surveys. Part II. Noise in Buildings.'' R. S.
Tucker. Noise experienced indoors is in one sense more important
than that experienced outdoors, for, with the growth of our industrial
civiUzation, increasing numbers of people are spending most of their
waking hours indoors. They are thus exposed to indoor noise for a
large part of the time, including the hours of work when noise has its
opportunity to impair their working efificiency.
Certain typical values for noise in various locations in buildings
have been published, and are summarized in this paper. Our knowl-
edge of indoor noise levels is far from complete, however. Further
information has been obtained in a survey of room noise in New York
City and the surrounding area which was made in 1929 by the National
Electric Light Association and the American Telephone and Tele-
graph Company in the course of the work of their Joint Subcommittee
on Development and Research. Some results of the New York City
measurements are given. About 70 test locations are included. It
will be realized that this is only a small sample of the total number of
places where indoor noise is experienced in New York City alone. The
conclusions given must therefore be regarded only as suggestive rather
than as holding true in any general sense.
' Jour. Acous. Soc. America, July, 1930.
Contributors to this Issue
Charles B. Aiken, B.S., Tulane University, 1923; M.S. in Electrical
Communication Engineering, Harvard University, 1924; M.A. in
Physics, 1925. Bell Telephone Laboratories, 1928-. Mr. Aiken has
been engaged on work in connection with aircraft communication and
more recently with the design of broadcast radio receiver equipment.
F. E. Haworth, A.B., University of Oregon, 1924; M.A., Columbia
University, 1929 ; Bell Telephone Laboratories, 1925-. Mr. Haworth's
work has been in crystal analysis by means of X-rays, magnetic mate-
rials, and more recently in studies of dielectrics.
Herbert E. Ives, B.S., University of Pennsylvania, 1905; Ph.D.,
Johns Hopkins, 1908; assistant and assistant physicist. Bureau of
Standards, 1908-09; physicist, Nela Research Laboratory, Cleveland,
1909-12; physicist. United Gas Improvement Company, Philadelphia,
1912-18; U. S. Army Air Service, 1918-19; research engineer, Western
Electric Company and Bell Telephone Laboratories, 1919 to date.
Dr. Ives' work has had to do principally with the production, measure-
ment and utilization of light.
W. C. Jones, B.S. in E.E., Colorado College, 1913; Western Elec-
tric Company, 1913-25; Bell Telephone Laboratories, 1925-. As
Transmission Instruments Development Engineer, Mr. Jones has
specialized in the development and application of instruments for the
transmission of speech and music.
A. R. Kemp, B.S., California Institute of Technology, 1917, M.S.,
1918; Engineering Department, Western Electric Company, 1918-25;
Bell Telephone Laboratories, 1925-. Mr. Kemp has been engaged in
chemical research on rubber and allied materials used for submarine
and other types of insulation.
W. H. Martin, A.B., Johns Hopkins University, 1909; B.S., Mas-
sachusetts Institute of Technology, 1911; American Telephone and
Telegraph Company, Engineering Department, 1911-19; Department
of Development and Research, 1919-. As Local Transmission En-
gineer, Mr. Martin has been engaged in development work on the
transmission of telephone sets and local exchange circuits, transmission
quality and loading.
153
154 BELL SYSTEM TECHNICAL JOURNAL
L. J. SiviAN, A.B., Cornell University, 1916; Engineering Depart-
ment, Western Electric Company, 1917-19 and 1920-25; Bell
Telephone Laboratories, 1925-. Mr. Sivian's work is in acoustics,
chiefly in connection with methods of electroacoustic measurements.
George C. Southworth, B.S., Grove City College, 1914, M.S.,
1916; Ph.D., Yale, 1923; assistant physicist. Bureau of Standards,
1917-18; instructor, Yale University, 1918-23; Information De-
partment, American Telephone and Telegraph Company, 1923-24;
Department of Development and Research, 1924-. Mr. South-
worth's work in the Bell System has been concerned chiefly with
the development of short-wave radiotelephony. He is the author of
several papers on radio-frequency phenomena.
The Bell System Technical Journal
April, 1931
Symposium on Coordination of Power
and Telephone Plant*
Introductory Remarks
By R. F. PACK
I UNDERSTAND I am expected to outline shortly what has led to
the splendid cooperation between the Associated Companies of the
American Bell Telephone System and the Power Companies of the
United States in the matter of coordinating their facilities to avoid
interference with the service of either.
Previous to 1921 disputes of a very serious nature were constantly
occurring between the Bell Telephone Companies and the Power
Companies, the former claiming that the rapid construction of trans-
mission lines by the latter was seriously interfering with telephone
service. The Power Companies felt that they also had a duty to
serve the public and resented the attempts of the Bell Companies to
interfere with the Power Companies' growth and progress. These
disputes were so acrimonious and the parties to them so bitterly dis-
posed towards each other that the courts and public service commis-
sions in the various states were more and more frequently called upon
to adjudicate the differences.
In the latter part of 1920 it was evident that the situation was be-
coming a serious menace to both great interests and suggestions were
forthcoming from certain individuals representing both interests that
attempts should be made to find a solution. Unfortunately, the names
of those responsible for this constructive thought are not known and
they cannot, therefore, personally be given their due meed of praise,
nor assigned their proper places in history. However, as a result, early
in 1921 a group of power men met with a group of Bell Telephone men,
under the neutral chairmanship of Mr. Owen D. Young and there was
then formed a permanent committee which has since been known as the
Joint General Committee of the National Electric Light Association
and the Bell Telephone System.
* Joint work of the National Electric Light Association and Bell Telephone
System. Presented at the Winter Convention of the A. I. E. E., \e\v York, X. V.,
January 26-30, 1931.
155
156 BELL SYSTEM TECHNICAL JOURNAL
This General Committee asked Mr. Bancroft Gherardi, Vice Presi-
dent of the American Telephone and Telegraph Company, and myself
to select a Subcommittee of Engineers representing both interests,
whose duty it should be to classify the types of physical situations in
which engineering or technical conflicts were arising between the two
interests and to indicate how on the basis of the existing state of the art
the electric light and power engineers considered such situations should
be met from a physical standpoint and how the telephone engineers con-
sidered such situations should be met without regard to the question
of division of costs.
We requested this Subcommittee of Engineers to approach the vari-
ous problems outlined in the broadest possible spirit of cooperation
bearing in mind that the object to be attained was the removal of
friction and the early development of mutually satisfactory standards.
Nearly a year later, in March 1922, Mr. Gherardi and I made our
first report to the Joint General Committee based on the conclusions of
the Subcommittee of Engineers.
Certain general statements were agreed to as for instance that the
National Electrical Safety Code provided an acceptable guide to prac-
tise and that there were substantial advantages to both utilities in the
employment of jointly occupied poles where conditions and character
of the circuits permitted. It was also recognized that the public's in-
terest was paramount and that both the power and communication
utilities must be able to render their respective services to the public
in an economical and efficient manner. A few general principles for
the solution of inductive interference situations were suggested such as
cooperative planning of all new construction and the further recom-
mendation that standards of construction and operation in accord with
the general principles outlined should be prepared and agreed to by
further cooperative work of the Subcommittee of Engineers, and finally
that a cooperative study of the art should be made in order to determine
what practicable measures, if any, might be developed and adopted to
lessen the contributing characteristics of both systems in this matter of
inductive interference.
Mr. Gherardi and I in reporting to the Joint General Committee
stated we believed great progress had been made and we urged that the
General Committee advise the power companies and the associated
companies of the Bell System to use every efYort to arrive at a settle-
ment of their differences through negotiations rather than resort to
court or commission proceedings. It will be noted here that after one
year we had made apparently but little progress in the actual solution
of the problems involved. As a matter of fact, we know now that the
INTRODUCTORY REMARKS 157
foundation stone had then been well and truly laid. It was not so
much what had actually been accomplished that mattered but that
the whole spirit of the relations between the telephone and power inter-
ests had been completely changed from one of friction, distrust, sus-
picion and even of enmity to one of confidence, good will and a desire
on the part of both to cooperate.
From that time the work progressed much more rapidly and in
December 1922 a reasonably complete set of principles and practises
for the inductive coordination of power and telephone systems had been
agreed to and sent to the member companies of the N. E. L. A. and
the associated companies of the Bell System over the signature of the
Joint General Committee of which, as I have stated, Mr. Owen D.
Young is Chairman. Since that time further reports containing prin-
ciples and practises for the joint use of wood poles and the allocation
of costs of coordinative measures have been agreed to and promulgated
by the Joint General Committee.
Today inductive coordination as between the Bell Telephone System
and the power companies is no longer a problem but only a routine
day to day job of cooperatively continuing research work and develop-
ing the art of both systems to eliminate as far as possible causes for
inductive interference.
I remember Mr. Gherardi once made the statement that the term
"problem" is generally applied to a thing where you do not know the
answer — "job" where you do know the answer to it and it is just a
question of working on it — and it is exactly at that point we have
arrived today. I do not mean to say we can remain quiescent as to
this work because it is still a big job and will require the attention of the
executives of the companies concerned and the constant and concen-
trated effort of the engineers of both interests who are engaged in
research and other necessary work connected with inductive coordina-
tion.
To have had some part in bringing about these results has been one of
the most satisfactory things I have done in my entire life and I believe
Mr. Gherardi will fully coincide with this viewpoint as far as he is
concerned. From the time I first met him, we have never departed
from our belief that the problem could be solved on the basis of entire
confidence, good faith and complete cooperation.
In the first instance we had many disappointments and some difficult
situations to combat but I can truly say that we never had a serious
disagreement and always were confident that the goal we desired would
eventually be reached. I remember making a statement in those
early days that I did not believe that each utility had obtained every-
158 BRLL SYSTEM TECHNICAL JOURNAL
thing that each utility wanted but that I was confident that both util-
ities had got what both utilities wanted, and that a problem of this
kind could not be settled by one party to a dispute getting all its own
way because then nothing was settled. The trouble would simply be
aggravated, making it more possible for controversies to arise again and
again. I added that at no time had there been any question of com-
promising on principles, nor bargaining across a table, — we have had
always before us a clear recognition of the problem of the other side
and a mutual admission of the fact that the other system must live
and that the primary interest is the public's and that the public must
efficiently and economically be served by both utilities.
It may be of interest to you to know that the power companies
with the same personnel on a General Committee, also headed by Mr.
Owen D. Young, are now carrying on similar cooperative work with the
Western Union Telegraph Company and with the Railroads with re-
spect to their signal systems. The result of our cooperative work with
the Western Union Telegraph Company will, of course, favorably affect
our relations with the other telegraph companies of the country, as our
work with the Bell System has affected in a highly satisfactory way
our relations with the independent telephone companies of the country.
May I in conclusion thank you for the privilege of making this
statement. It has been a particular pleasure to me because I am more
and more convinced that this is the sound way to settle such problems
and countroversies arising between great interests in this country.
Courts and regulating authorities approve this method because it
promotes harmony and permits them to devote their time and talents
to other useful purposes and because it saves the taxpayers the material
expense of costly technical hearings in which the interests of the public
are in no way jeopardized.
Trends in Telephone and Power Practise as Affecting Coordination
By W. H. HARRISON and A. E. SILVER
The general trends in telephone and electric power systems are ontlineil
and the reactions of certain of these trends on coordination are described.
In the telephone system, brief mention is made of the rapid growth of
the dial system of operation, improvements in subscriber-station apparatus,
rapid extension of new types of facilities for toll circuits and the growth of
connections to foreign countries. Improvements in telephone service
increase the importance of securing adequate coordination. The advantages
of the use of cable facilities for toil circuits, of repeaters, of carrier current
systems as regards coordination of long distance and interurban telephone
circuits are discussed. The benefits accruing from improved subscriber-
station apparatus, central office equipment, abandonment of iron wire for
the short tributary toll circuits and new methods of making sleeves at joints
in open wire lines are outlined.
In the power system, brief mention is made of increasing use of larger
generating units, and growing use of automatic devices to replace manual
operation. Improvements in power service generally react favorably on
coordination. The general trends toward higher voltages for transmission
and distribution and the improved standards of construction accompanying
these trends are described. The important matter of system stability and
the practises as regards grounding of transmission circuit neutrals, lightning
control and current limiting devices, and the reactions of these matters on co-
ordination are outlined. Reference is also made to grounding of distrilaution
svstem neutrals, service taps on transmission lines, general practises as
regards transformer connections and improvements in wave shape in so far
as these matters react on coordination.
In conclusion, it is pointed out that, while there have been mfluences
working both favorably and unfavorably toward coordination, the pre-
ponderant trend is defin'itelv toward an improvement. The benefits which
have accrued from the activities of the Joint General Committee and the
important function of the Joint Subcommittee on Development and Re-
search are also mentioned.
T
General Trends
â– HE important benefits resulting from the cooperative handUng of
questions arising from the proximity of the physical plants of the
telephone system and the electric power systems of the United States
are emphasized when consideration is given to the e.xtent and the rapid
growth of these two industries. This growth is illustrated by Fig. 1
which shows that during the past decade, while the population of the
country has increased 16 per cent annual telephone messages have in-
creased 96 per cent and annual kilowatt hour usage of power 107 per
cent. Another indication of the grow^th of these utilities is given by
Fig. 2 which shows that during the past decade customers telephone
* Part I of the Symposium on Coordination of Power and Telephone Plant.
Presented at the Winter Convention of the A. I. E. E., New York. X. V., January
26-30, 1931. Published in abridged form in Electrical Engineering, March, 1931.
159
160
BELL SYSTEM TECHNICAL JOURNAL
stations have increased 88 per cent and customers of central stations 127
per cent. The leaders of both utilities confidently expect that, apart
from temporary setbacks a.ssociated with recessions in general business,
the recent rapid growth of these utilities will continue throughout the
next decade.
- 80
O
<
Q
(0
o
CL
in
Z
2
o
Irt
iij
a.
Z
D
1
1930 Population 123,900,000
Telephone messages 21,600,000,000
Kw. hours generated 90,000,000,000
Fig. 1 — Per cent increase, 1920 to 1930, in population and in telephone and
power usage.
Note: Values for 1930 are estimates based on best available data. Telephone
data refer to Bell System.
Such a rapid growth of the two utilities both of which must supply
the same customers with services essential to their comfort and pros-
perity, necessarily brings with it a large number of cases of physical
proximity between the plants of the two utilities where, due to the
widely different characteristics of the circuits involved, difficulties may
arise. The necessity for active study of the coordination of the differ-
ent systems and for the current handling of large numbers of individual
situations will continue for a long time to come.
Associated with this rapid growth there has been another trend in
these two utilities which has an important effect on coordination work.
TRENDS IN PRACTISE AS AFFECTING COORDINATION 161
This trend is the steady improvement in the quaHty of service afforded
to their customers.
In the telephone system the improvement in the standards of service,
if considered by itself, tends to increase the noticeability and the re-
action on service of inductive effects from outside sources. Such
changes as the improvement in the characteristics of transmitted
speech, including the extension of the band of frequencies efficiently
transmitted, and the avoidance of cases in which interfering noises
are produced from sources within the telephone plant, tend to increase
9 IS
/
POWEP CUSTO^
EPS
/
^
.^---^fELEPHON
E STATIONS
^^
Fig. 2 — Telephone station and power customer growth.
Note: Values for 1930 are estimates based on best available data,
data refer to Bell System.
Telephone
the effect of moderate amounts of noise current induced in the tele-
phone circuits from outside sources. Similarly increases in the extent
of the service and in the speed of completing calls have led to increased
reliance on prompt telephone communication which tends to increase
the importance of avoiding interruptions. Five years ago the average
interval of time between the placing of a long-distance toll call by a
subscriber and the commencing of the conversation was 1)4 minutes.
At the present time it is a little less than 2>^ minutes. Telephone
users have now come to rely on the almost immediate establishment of
telephone connections and are correspondingly more critical of inter-
ruptions or delays.
162
BELL SYSTEM TECHNICAL JOURNAL
The improvement of service has been associated with a particularly-
rapid growth of very long haul telephone business and a consequent
increase in the average length of telephone circuits used for interurban
and long distance work. This is illustrated by Fig. 3 which shows the
160
120
O 100
o
a. 80
ui
2
60
40
20
/
^ /
/
/
.^
^
NEW YORK-/
CHICAGO /
^ y
^^OS
TON - NE
.W YORK
/
/
/
^
/
/
NEW YOR
CHICA
TO
^ AND
30
— -
^^
/ «
5AN FRAN
go LOS Ar
CISCO
JGELES
1920 1922 1924 1926 1928 1930 1932 1934
YEAR
Fig. 3 — Long haul telephone circuit growth of typical circuit groups.
growth in the last few years and the expected growth for the next few
years of typical circuit groups of different lengths. In the period 1925
to 1929 while telephone toll business as a whole increased 59 per cent
New York-Chicago business increased 170 per cent and the combined
Chicago and New York business to Los Angeles and San Francisco
380 per cent. From the standpoint of coordination with other electric
circuits the very long telephone circuit offers a more dif^cult problem
than the circuit of moderate length because of the cumulative eflect of
exposures in different sections.
TREXnS IX /'RACTISK AS AFFECTING COORDIXATIOX 163
In the power industry one of the most important items in the im-
provement of service has been the stead>' decrease in the number of
service interruptions. This has been brought about mainly by better
standards of construction, including more systematic mechanical and
electrical arrangements of circuits and apparatus, and increased num-
bers of circuits and sources of supply. The interconnection of power
systems has figured largely in the last mentioned factor contributing to
service reliability, by making available greater numbers of sources and
by multiplying the routes over which power can be recieved at specific
locations. While the increasing numbers of interconnecting and other
types of lines bring new conditions for the coordination of power and
telephone plants, improved construction and increased security of
circuits and apparatus have a definitely beneficial effect upon matters
of coordination by reducing the number of abnormal conditions of
operation.
Other items in the improvement of the service given by the power in-
dustry are better voltage regulation and a great increase in the number
of types of power consuming appliances and apparatus made available
for the customer. Accompanying better voltage regulation are certain
factors which definitely aid coordination, among these being better
balance of currents in the separate phases of the circuits and more
effective arrangements minimizing the tendency for currents to flow in
the earth. The effect of increased numbers of types of utilization
apparatus on coordination is problematical, though probably not of
sufficient magnitude to be of practical importance.
Other trends which have a bearing on the improvement of power
service are discussed in the section of this paper devoted to the power
system.
While in some respects the general trends indicated above, namely,
the extent and rapid growth of the two utilities, and the improvement
of service standards, have by themselves tended to increase the im-
portance and the difficulties of coordination work, these adverse ten-
dencies have been offset by beneficial effects of improvements in plant
design and construction and by the cooperative endeavor which has
been carried on by the two utilities during recent years. It is a tribute
to the effectiveness of this cooperative work that the degree of satis-
factory coordination between the two systems is steadily improving.
Fig. 4 shows that during the past 10 years the mileage of telephone
toll circuits has increased 250 per cent and the mileage of power trans-
mission lines over 100 per cent. The effect of such growth on the num-
ber of situations of proximity is illustrated by the fact that during the
past three years the exposures of interest from a noise standpoint have
164
BELL SYSTEM TECHNICAL JOURNAL
increased from the equivalent of about 10 miles to about 14^ miles per
100 miles of open-wire telephone toll lead; while on the other hand the
exposures not as yet adequately coordinated have in the same period
decreased from the equivalent of 2.6 miles to 1.5 miles per 100.
t
/
/
/
/
/
TOLL
TELEPHONE y
CIRCUITS/^
'
y
^^
^
— ■—
POWER
TRANSMISSION
CIRCUITS
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ifi Q
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^8
(El
UJl_
Fig. 4 — Toll telephone and power transmission circuit growth.
Note: Values for 1930 are estimates based on best available data. Telephone
data refer to Bell System.
While the trends of practise in the design, construction and main-
tenance of the plants have necessarily been largely controlled by the
fundamental requirements of service and economy in developing the
two systems, and while the trends naturally have not all been in the
same direction as regards their effect on the coordination problem, still
the general trend of plant practise at the present time is in the direction
to facilitate the coordination of the plants of the two utilities. In the
following pages brief statements are made, descriptive of the more im-
portant of these trends in the respective systems.
Trends in Telephone System
The telephone plant is at the present time rapidly changing in its
physical character through the application of important developments
and changes in engineering and construction practise.
TRENDS IN PRACTISE AS AFFECTING COORDINATION 165
Probably the most fundamental and far reacliiiii; of these changes is
the progress of conversion from manual to dial system operation.
When present plans are completed this will result in the operation of
approximately 80 per cent of the telephones of the Bell System on a
dial basis, and a large part of the existing manual central ofifice equip-
ment will have been removed from service. \Mth the application of
the dial system there is a trend toward a greater concentration of cen-
tral office equipment in one building, so that in the future as many as
100,000 telephones may be switched by the various central office units
in a single building. While these trends are of the greatest and most
fundamental importance from the standpoint of the development of
the telephone business they do not atTect the coordination problem in
any material way and therefore need not be further discussed here.
An important trend in telephone practise has been the provision of
apparatus designed for higher standards of service and greater con-
venience for use at the customer's station. This includes the hand set,
new types of private branch exchanges and of auxiliary telephone
station apparatus, and improvements of transmission characteristics.
These changes in some respects affect the coordination problem and
these effects are indicated below.
Another important fundamental change in the telephone plant and
one of great importance from the coordination standpoint is the rapid
extension of new types of facilities for toll circuits, that is, long distance
and interurban circuits whose use involves what is called a toll charge.
These changes and their effects on the coordination problem are dis-
cussed in this paper.
One of the most spectacular trends of development of the Bell System
at the present time is the increase in the number of connections to
foreign countries. Earlier connections to Canada and Cuba were
supplemented in 1927 by service to Mexico and by transoceanic radio
links providing service from New York to London, through which
connection is made to the principal European countries; and in 1930 a
similar radio link from New York to Buenos Aires through which con-
nection is made to Montevideo, Uruguay, and Santiago, Chile. Dur-
ing the next few years it is expected that these foreign connections will
increase to include generally all important points in South Amenca,
Australia, Japan, Honolulu and all other points which may offer an
appreciable demand for service.
These intercontinental circuits are not of such character and location
as to be directly affected by the physical proximity of power circuits,
but their efficiency is affected by the noise currents on connected cir-
cuits in the same way as other very long circuits are affected and this is
discussed briefly below.
166
BELL SYSTEM TECHNICAL JOURNAL
Toll Cable. — The change in methods c^f designuig and constructing
toll circuits which is of greatest importance from the standpoint of
general development of telephone plant is the great increase in use of
cables for those circuits, including both the very long distance circuits
and the sliorter interurban circuits. This increase is shown by Fig. 5.
o 5
in
z
o
Z 3
1
1
1
1
1
1
/
1
1
/
/c^Ql
E
/
Y
^
'^^
—
OPEN
WIRE
^
CARF
1
lER
>«•
1921 1922 1923 1924 1925 1926 1927 1928 1929 1930
YEAR
Fig. 5^Toll telephone circuit growth by classifications.
A single cable may provide for from 250 to 500 telephone circuits and
several hundred telegraph circuits, that is, as many circuits as would be
provided by five to ten heavily loaded pole lines of aerial wire construc-
tion. This concentration of circuits in a single cable, a number of
which can be placed on a single route, is in itself of great assistance in
coordination problems by greatly reducing the number of routes for
which coordination arrangement nmst be made, l^^urthermore, the
TREXDS IX PRACTISE AS AFFECTI.XG COORDIXATIOX 167
presence of the lead sheath, together with the twisting of the cable
conductors, the high degree of balance with respect to ground, and the
mutual shielding effect of the many circuits in one cable, practically
prevents noise currents from being induced directly into the cable cir-
cuits from outside electrical sources. The shielding etTect of the lead
sheath when suitably grounded also provides substantial reductions in
the voltages of fundamental frequency which may be induced along
the cable conductors at times of trouble on neighboring power systems.
A telephone toll cable with its associated equipment costs about the
same per mile as a twin circuit power transmission line of the 110-kv.
class. This high cost has led to a large use of private right of way for
new^ extensions of these cables, particularly for aerial cable construction.
This, of course, has an added advantage from the coordination stand-
point in tending to keep these important telephone routes off the high-
ways, which are so much used for the distribution systems of both
utilities. In the more rapidly growing cable routes underground con-
duit construction is employed and these in most cases are located along
the highways. In these cases, however, the close proximity of several
cables in the same conduit run offers a considerable amount of mutual
shielding effect which reduces the susceptiveness of circuits in these
cables to values approaching that obtainable by a single tape armored
cable.
This tape armored cable, w^hich recently has been placed in use in
this country, is designed for burying directly in the ground, and has an
increased degree of magnetic shielding. This is provided by two wrap-
pings of steel tape outside the lead sheath which are necessary for the
mechanical protection of the cable when ducts are not used. During
the past year about 160 miles of this cable were installed and it is ex-
pected to have a considerable field of use in the future.
As indicated above, in all these types of cable construction the sus-
ceptiveness to noise induction is so greatly reduced that low frequency
induction generally becomes the limiting factor relative to the permis-
sible proximity of these cables to power circuits. The relative amounts
of induced voltages with these different types of construction in com-
parison with open wire construction, while naturally varying with local
conditions, are indicated in a general way in Table I.
TABLE I
Approximate Relative Volts on
Telephone Circuits per Ampere
of Inducint; Current at
Type of Construction 60 Cycles
Open wire ^ -^
Single cable, aerial or underground — sheath well grounded O.o
Buried tape armored cable — well grounded U--
Note: All values for cables assume full size, i.e., 2;\s-in. diameter.
168 BFXL SYSTEM TECHNICAL JOURNAL
The above figures are based on favorable conditions for obtaining
low resistance ground connections on the cable sheaths. Such ground
connections are necessary to provide the full shielding benefits, since the
shielding is brought about by induced currents on the cable sheath
flowing along the sheath and through ground. These sheath currents,
because of the close coupling between the sheath and pairs, induce
voltages into the pairs tending to neutralize the voltages induced into
the pairs directly from the power system. The use of the tape armor,
which is a magnetic material, increases the coupling between the sheath
and pairs. The grounding conditions necessary for satisfactory shield-
ing effects can usually be obtained, but situations sometimes arise in
the case of aerial construction where it is difficult or impossible to
obtain them.
While as noted above, the cable circuits are effectively protected
from noise induction, the efficiency obtainable over the long circuits is
limited in part by the noise currents occurring in the open-wire lines
which may be switched to the long cable circuits. This is because the
efficiency of the long cable circuits depends upon voice-operated
switching devices which must not be operated by the noise currents.
This is also true of the intercontinental circuits mentioned above. The
extension of the circuits controlled by voice-operated devices tends
therefore to increase the importance of good coordination of the entire
plant.
Telephone Repeaters. — Another important trend of practise is the
extended use of telephone repeaters. The purpose of these devices is
to amplify the voice currents and thus make possible higher efficiency
and greater extension of long distance telephone circuits. Their use
is essential to the great development of toll cable. Moreover, they are
used widely on open-wire circuits. Without repeaters it was necessary
on the long open wire circuits to permit the power level of voice currents
to sink to relatively low values. An extreme example of this is given
by the New York-Denver circuit which, before repeaters were available
for use on this circuit, had an overall equivalent, using the highest
grade of telephone construction which had been developed up to that
time, of about 31 dhJ With the application of repeaters to this circuit
the level of voice currents could be kept relatively high throughout the
circuit. This is illustrated in Fig. 6 giving level diagrams for the cir-
cuit as originally set up and later when provided with repeaters.
The use of repeaters contributes to reducing the susceptiveness of
the telephone plant and thus aids coordination. On such a circuit as
the original New York-Denver circuit just mentioned, a relatively
' This means that the ratio of output power to input power of this circuit is O.OOOS.
TRENDS IN PRACTISE AS AFFECTING COORDINATION 169
small amount of noise current greatly impaired transmission because
of the weak incoming voice currents. Although the repeaters naturally
amplify the noise currents as well as the voice currents, the fact that
the voice level is kept high throughout results in great benefit which in
this case, assuming similar exposure conditions in the various repeater
sections, gives an improvement in the ratio of voice currents to noise
currents of slightly over five.
1000
DISTANCE IN MILES
Fig. 6 — New York^Denver circuit level diagrams.
Repeaters probably also have some effect in reducing certain of the
eft'ects of low frequency induction by the fact that they sectionalize
cable lines at about 50 mile intervals and open-wire lines at intervals of
200 miles or less, and limit the power which can be transmitted from
section to section. There is some evidence that this tends to limit
acoustic shocks.
Carrier Telephone Systems. — A third important trend in telephone
practise is the extension in the use of carrier telephone systems for
long circuits and the associated changes in aerial wire construction
practises. The growth in use of this type of circuit is indicated in
170
BELL SYSTEM TECIIMCAL JOIRXAL
\'\'g. 5. The carrier systems are much less iiiHuenced by noise iiuluc-
lion from power circuits because they occupy a range of frequencies
(5000 to 30,000 cycles) in which the harmonic power voltages or cur-
rents ordinarily are extremely small. Furthermore, in order to obtain
economies inherent in the use of large numbers of carrier systems on the
same telephone pole line it has been necessary to design systems of
3-C
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8v8 6v8 I
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V
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T
BH
VOICE FREQUENCY-PHYSICAL
VOICE FREQUENCY-PHANTOM
CARRIER TELEPHONE
D-C TELEGRAPH
CARRIER TELEGRAPH (10 CHANNELS)
TOTAL TELEPHONE
TOTAL TELEGRAPH
TOTAL CIRCUITS
20
2
48
40
40
70
80
Fig. 7 — Pole line configuration.
Xon-piiantonied construction — 8-inch spacing between wires of non-pole pairs.
transpositions of much increased effectiveness and even to change the
configuration of the wires in order to greatly reduce the inductive effects
between the telephone circuits. These changes also result in reduced
susceptiveness to outside inductive influences. The type of construc-
tion now recommended for new aerial wire lines in cases where the
extensive use of carrier is anticipated is shown in Fig. 7. The two
wires of each pair, except pole pairs, are spaced 8 in. apart compared
TRENDS IN PRACTISE AS AFFECTING COORDINATION 171
with the previous standard of 12 in. Often transpositions are made
as frequently as every second pole and are of an improved type givin.g:
better balance between circuits; also on the circuits on which carrier
telephone is used the i:>hantoms are abandoned. The relative suscep-
tiveness to noise frequency induction of the various types of aerial
wire construction has been tested for various typical conditions. The
results of these investigations are summarized in Table II.
TABLE II
Type of .Approximate Relative
F'lcility Transposition* Susceptiveness t
1 2 in. phantom Voice (brackets) 1.00
12 in. side Voice (brackets) 0.50
8 in. pair Carrier (break irons) 0.25 or less
* Voice circuits are not so frequently transposed as carrier circuits. Bracket tvpe
transpositions require two spans to complete the transposing whereas the break iron
type completes the transposing on a single crossarm.
t Susceptiveness is used in the sense defined by the Joint General Committee,
namely, "Those characteristics of a signal circuit with its associated apparatus which
determine, so far as such characteristics can determine, the extent to which it is
capable of being adversely affected in giving service, by a given inductive field."
Subscribers' Station Apparatus. — To a large extent the trend of
development in subscribers' station apparatus is toward new arrange-
ments which provide greater convenience and more closely meet the
needs of the users and which have no material effect upon the coordina-
tion problem. An important group of developments, however, centers
about the improvement of the electrical performance of the station
apparatus by removing impairments caused by the earlier types of
apparatus. These changes, by improving the quality of speech as
reproduced by the telephone system, tend to make more noticeable
the impairments caused by the effects of currents induced from external
sources.
The tendency toward an increase in the range of voice frequencies
efficiently reproduced by the telephone system tends to increase the
range of frequencies of induced currents which may cause noise inter-
ference as discussed in the introductory section. An extreme illustra-
tion of this is the circuits designed to transmit programs for radio
broadcasting stations. The transmission characteristics of these cir-
cuits have been improved by including both higher and lower fre-
quencies, and in their most modern form these circuits efficiently trans-
mit currents of frequencies in the range between 35 cycles and 8000
cycles and are therefore capable of being affected by inductive noises
over this wide range.
The room noise conditions at the subscribers' premises have an effect
on telephone transmission. This noise besides acting directly on the
172 BELL SYSTEM TECHNICAL JOURNAL
ears of the telephone user is converted by the transmitter into electrical
currents, a part of which actuates the receiver, thus producing noise.
The present trend in telephone practise is very strongly toward a
reduction of these effects. This will tend to bring into increasing
prominence noise caused by induction in the telephone circuits which
now in many cases is partially overshadowed by the reproduction of
the noises in the room.
As partly offsetting this tendency steps have been taken to improve
the degree of balance to ground of new station apparatus, particularly
in the case of party lines. The new station apparatus with the
improved transmission characteristics discussed above will be designed
for reduced effect of noise currents entering from the line. Also, in
extending the selective signaling features to rural areas, higher im-
pedance ringers and a newly developed high impedance relay are being
used in order to limit susceptiveness to noise from exposures between
the rural open wire extensions and rural power distribution circuits.
Where central office equipment is being modified to permit of increased
range of direct current signaling, or for some other reason, the reduction
of susceptiveness is always a consideration. All of the newer repeating
coils used for supplying talking battery to subscribers in common
battery areas, which comprise the bulk of the local plant, possess a
much higher degree of balance than the coils which were standard a
few years ago.
Other Items. — So far the changes which are associated directly with
the major trends of development in the telephone plant have been
described. The broad outlines of these developments depend on all of
the factors affecting telephone service as well as coordination with
power circuits. There are other features not directly associated with
these main trends which, while introduced into the telephone plant
largely because of the advantages to be gained in reducing susceptive-
ness to electrical influences, have also afforded other benefits. A few
of the more interesting examples of these changes are given below.
Referring to the toll plant, there may be mentioned the recently
adopted general practise of soldering aerial wire sleeve connections in
order to insure a permanently high degree of series balance. Hereto-
fore reliance had been placed on the contact between the wires and the
twisted sleeve. The practise of soldering will be supplemented in the
near future by a cold-rolled sleeve method, and it is confidently ex-
pected that these practises will result in material noise improvements.
They will also probably reduce the maintenance required on open wire
toll circuits, particularly where exposures are involved.
Another item is the abandonment of the use of iron wire and sub-
TRENDS IN PRACTISE AS AFFECTING COORDINATION 173
stitution of copper for short tributary toll circuits. Coordination of
the iron wire circuits is relatively difficult because of the development
of resistance unbalances at the wire joints. The transmission efficiency
is also improved by the reduced resistance afforded by the copper but
this effect is generally of secondary importance in the short tributary
circuits.
In toll offices improvements have been made in the balance of coils
and condensers used for superposing telegraph on the telephone circuits.
The use of repeating coils, commonly used for side-circuits, has been
extended to phantom toll circuits. These coils act as insulating trans-
formers to prevent noise voltages from the outside conductors being
impressed upon the intricate cabling and equipment of the office.
Referring to the local plant, there are several noteworthy examples
of modifications made principally for the purpose of reducing suscep-
tiveness. Investigation such as that of the coordination between
power and telephone distribution plants conducted at Minneapolis
by the Joint General Committee, stimulated the development of means
for reducing the susceptiveness of the telephone distribution plant.
Present practises call for the interconnection of aerial and underground
cable sheaths and the grounding of the aerial sheath in order that the
benefits of the shielding action of the sheath currents as previously
described, may be realized for noise induction. In cases where elec-
trolysis conditions do not permit direct grounding, condensers of the
electrolytic type are employed to prevent the flow of direct currents.
The telephone circuits have long been equipped with over-voltage
protectors for the purpose of protecting apparatus and cables against
lightning waves and against power frequency transients from the lower
voltage distribution circuits, also with fuses for opening the lines in
cases in which heavy currents flow. The trend in development of these
devices has been principally toward more uniform operation and lower
maintenance costs. With the rapid increase of voltage and capacity
of power circuits generally, experimental studies have been undertaken
of further means for maintaining the safety of persons working on or
listening on the telephone circuits. At the present time, development
work is being done on various devices for this purpose, some of which
are fundamentally different in design and operation from those pre-
viously used. It is hoped that these devices, which are discussed in
one of the following papers, will afford increased protection against
overvoltages and improve coordination conditions.
Trends in Power System
In the field of power generation marked attention has been paid,
from the start, to methods of improving the efficiency of the generating
174 BULL SYSTEM TECIIXICAL JOCRXAL
process and reducing the investment per kilowatt of generating ca-
pacity. This has led to the development of larger and larger generat-
ing units. A single shaft unit of 160,000 kw. capacity and a triple-
element unit of 208,000 kw. capacity are in operation. The latter
consists of one high pressure and two low pressure turbines with their
respective generators. Single shaft units of 200,000 kw. capacity are
under construction and it seems probable that the trend in the future
will be toward even larger units of both types. This trend toward
larger units instead of the equivalent in small units has resulted in
improved wave shape but otherwise does not directly affect coordina-
tion except in so far as it may reflect the general trend toward larger
concentrations of power with the accompanying tendency to increased
magnitude of system abnormals.
Another definite trend in the power industry, but one which is not
of importance from the standpoint of coordination, is the increasing
use of automatic devices to replace manual operation. Complete
automatic operation is being practised to some extent in hydroelectric
generating stations and is widely practised in substations of various
types. The trend is definitely toward wider use of automatic devices
and new types and applications of such devices are being constantly
developed.
In view of the remarkable development and rapidly multiplying uses
of thermionic tubes and related devices in other fields, and the theoreti-
cally potential applications in the power art, the question will doubtless
be asked as to the trend of their application in the power field. How-
ever, other than application for current rectification, such as in railway
work, it cannot be said that progress has advanced to the point of
establishing a trend.
Those trends in power system development which are more directly
concerned with matters of coordination are discussed in the following.
System Voltages.— Referring to Table III, it is of interest to note
that the rate of increase of transmission line mileage, as a whole, is
lagging behind the rate of growth of both installed generator capacity
and electricity production. P'urthermore, mileages of the higher
transmission voltages, 220 kv., 132 kv., 110 kv. and particularly 66 kv.,
are growing at a faster rate than the group average. These compari-
sons reflect the increasing utilization of the higher voltages with the
greater circuit capacities they provide. As power industry growth
requires the handling of larger l)locks of power and as greater distances
between sources and markets are encountered, the development and
use of circuits and apparatus to transmit at voltages higher than the
220 kv. initiated in 1923 must be expected as an economic necessity.
TRENDS IN PRACTISE AS AFFECTING COORD I NATION 175
In the distribution field also, coincident with the development of
rural service, there has been a movement to higher voltages in primary
circuits, and indications point to the continuance of this trend in the
future. Due to the distances involved, voltages from 6600 to 13,200
(and even higher) have been used in rural work. In urban areas the
high load densities encountered in some districts require the handling
of large blocks of power in the primary circuits, and the lower primary
voltages have often been replaced by higher voltages for such conditions.
In addition to the greater capacities provided by the higher voltages,
possibilities of system simplification by combining rural and urban
systems and eliminating voltage transformations are of considerable
economic importance.
While at first glance the pronounced trend to higher transmission
and primary distribution voltages may appear to enhance the difficult-
ies of coordinating communication and power lines, certain factors
enter to offset this. As transmission voltages increase, line construc-
tion as a whole becomes more massive, greater clearances and wider
rights of way become necessary and construction costs per mile
rapidly rise. These greater space requirements weigh against the use of
highway locations and, together with the higher construction costs,
which make the shortest possible lengths desirable from an economic
viewpoint, frequently influence the selection of direct cross-country
private rights of way providing generally greater separation from
communication circuits in the same territory.
TABLE III
Total Circuit Miles of Transmission Lines.
Years 1926-1929 Inclusive
By Voltages,
Voltages
1926
1927
1928
1929
Per Cent of
Total
1929
Average
Annual
Increase
Per Cent
1926-1929
220,000
1,054
3,125
7,875
12,157
8,801
7,517
23,831
10,130
19,496'
8,072
28,223
1,257
3,343
8,661
15,212
9,257
8,492
24,706
10,429
18,441'
9,145
28,535
1,442
4,010
9,114
18,716
8,076
8,732
27,451
11,545
19,551
10,007
29,843
1,442
4,448
10,159
21,236
8,174
8,761
28,523
12,583
21,340
10,860
31,916
0.9
2.8
6.4
13.3
5.1
5.5
17.9
7.9
13.4
6.8
20.0
11.0
132,000
12.5
110,000
8.9
66,000
20.4
60,000
-2.4
44,000
5.2
33,000
6.2
22 000
i .:>
13 200
3.1*
1 1 000 ....
10.4
All other over 11, 000..-
4.2
Total
130,281
137,478
148,487
159,442
100.0
7.0
* This apparent discrepancy is believed to be due to reclassification of these lines
as between transmission and distribution facilities.
176 BELL SYSTEM TECHNICAL JOURNAL
The use of the higher voltage circuits, each transmitting many
thousands of kilowatts, of itself tends to increase the problems of
coordination. However, the greater separations obtained by the use
of private rights of way for these main transmission circuits in most
cases eliminate the need for coordinative measures to control normal
induction (manifested as noise in the telephone circuits) and, in case
noise presents a specific problem, the greater separations simplify and
render less extensive those specific coordinative measures which may be
required. Induction due to power system abnormals too is mitigated
or rendered easier of control.
In the case of distribution lines, the adoption of increasingly higher
voltages is accompanied by more systematic grades of construction
and greater clearances from communication circuits. The result, of
course, is that fewer abnormal conditions of operation occur and the
number of related disturbances in the communication circuits is cor-
respondingly reduced. The possibility of contact between power and
communication circuits is also reduced. This trend toward better
grades of construction applies also to transmission lines and, as noted
previously, to other parts of the power system.
System Stability. — During recent years considerable attention has
been paid to the development of methods for improving system electri-
cal stability. One of the most important of these methods is the use of
higher speed switching, — at present, faults can be cleared in 15 cycles,
or less, of a 60 cycle wave. So far, high speed switching has been ap-
plied mainly to transmission circuits. However, as development
proceeds and cost of equipment required is reduced, the field of appli-
cation of high speed switching may naturally be extended to distribu-
tion systems. The result in the case of either transmission or distri-
bution will be, of course, to reduce the duration of transients. Akin to
high speed switching, the use of high speed excitation of rotating equip-
ment has been developed. This may tend to increase the maximum
fault current values somewhat which would make coordination more
difftcult. However, the reduction in the severity of instability surges,
in so far as such surges involve faults-to-ground, affords definite bene-
fits from the coordination standpoint. It requires further study and
observations to determine what, if any, inherent limitations or advan-
tages it may possess with respect to coordination work.
The way has been paved for the development of high speed switching
by steady improvement in relaying practise. Selective operation of
protective relays in power systems, during the early stages of relay
development, was largely dependent upon an additive sequence of time
intervals which might aggregate a considerable period in the case of the
TRENDS IN PRACTISE AS AFFECTING COORDINATION 177
more remote units in the sequence. The development of relaying
practise has included various methods of securing selectivity independ-
ently of time. This has accomplished large increases in the over-all
speed of operation, at the same time improving selectivity. Coinci-
dent with these improvements there has also been a substantial gain
through greater precision in design and workmanship and improved
application of relays and related devices. These trends definitely aid
coordination by reducing duration of transients, eliminating faulty re-
lay operation, and steadily reducing the radius of influence of system
abnormals.
With the growth in power systems and major interconnections, the
use of bus or feeder current limiting reactors or other means of limiting
the concentration of fault current flow is being given increasing applica-
tion. Such practise acts to restrict the magnitude of inductive tran-
sients. In distribution systems the growing use of feeder reactors has
a similar effect in matters of coordination.
For well known reasons, among which are the avoidance of transient
over-voltages resulting from arcing grounds and the economies made
possible in apparatus insulation, it is predominant practise in America
to ground the neutrals of transmission systems at important trans-
forming centers, sometimes through resistors or reactors but usually
solidly. In view of the prevalence of the latter method, a large pro-
portion of higher voltage transformers now in service have been con-
structed with insulation between the neutral ends of the grounded
windings and the core and tank, designed to support only the neutral
potentials produced by fault currents regulating through the unavoid-
able impedance of grounding connections. The economies resulting
from this method of construction become greater as rated operating
voltages rise. The use of solidly grounded neutrals tends to make
coordination more difficult in view of the possibilities for increased
flow of earth currents.
On some large power networks with relatively great possible concen-
trations of short-circuit power and solidly grounded neutrals tenden-
cies towards instability of operation have appeared. In some instances
also oil circuit breaker characteristics, particularly as regards the older
breakers in service, have become a source of concern. For these rea-
sons, in these situations, increasing study and consideration are being
given to the use of current limiting devices in the neutral where the
characteristics of the apparatus and limitations of relaying will permit
of such operation.
In some European countries, particularly in Germany, where ground-
ing for the purpose of power system voltage stabilization is excluded
178 Hh.l.L SYSTEM TI-.CIINICAL JOURNAL
by governmental regulation, dependence is extensively placed on the
Petersen coil as a substitute. This device may be regarded as a special
type of neutral impedance. The Petersen coil has been applied to but
limited extent in this country although its possibilities for moderate
voltage systems, especially for situations warranting only single circuit
supply, are receiving consideration.
In this country, the increasing use of neutral impedance as well as
the use of other types of current limiting devices is an aid to coordina-
tion since it reduces the magnitude of abnormal induction.
Lightning Control. — The major problem of the transmission art at the
present time is the control of lightning in its effects on service. In
those sections of the country in which lightning is prevalent, this
natural hazard accounts for a large proportion of transmission circuit
faults, approaching 100 per cent in the case of the heavier, higher class
trunk transmission lines. The seriousness of this problem and the
researches which some of the larger power utilities and apparatus
manufacturers are conducting for its solution are being fully reported
from time to time before the Institute and need not be discussed here.
It is suflticient to say there is encouragement that methods for the solu-
tion of this problem, as it affects high voltage trunk circuits, will be
known in the not too distant future. Where adequate methods are
found and applied the results, of course, will be a decrease in the num-
ber of system disturbances which induce transients in communication
circuits.
Present measures in power system practise, especially at the higher
voltages, directed toward the control of service interruptions caused
by lightning include improved application of overhead ground wires,
improved grounding connections at the supporting structures, the
improved use of wood for lightning insulation, and the use in shunt with
line insulators of fused gaps or other valve devices to "spill" the surge
without dynamic current follow up. There is also under consideration
the application on grounded neutral systems of single-phase switching.
All of these measures, with the exception of the last, are helpful from
the coordination viewpoint since their effect is to avoid or reduce sys-
tem faults or at least to decrease the magnitude of earth fault currents
and hence of the accompanying voltages induced in nearby communica-
tion circuits.
Single-phase switching involves the use of individually controlled
and operated single-phase circuit breakers. Upon the occurrence of a
single-phase fault-to-ground, the breakers on the faulty phase only
would open, leaving the other two-phase conductors in circuit to main-
tain connection momentarily between source and load. In a short
TRE.ynS I.\ PRACTISE AS AFFF.CTING COORDINATION 179
inten-al the breakers controllini;- the faulty pliase would he reclosed
automatically.
Single-phase switching has not progressed beyond a preliminary
consideration of its possibilities. If applied in situations of proximity,
the residual voltages and load currents while one phase of a three-
phase grounded neutral system is momentarily open circuited may con-
stitute a problem in coordination.
Under 'ground Construction. — The use of underground construction
in distribution systems is seldom economical but is increasing in high
load density districts and in some residential areas primarily due to
requirements for civic improvements and the relieving of surface con-
gestion. The reduced influence on communication circuits of such
underground circuits as compared to overhead construction, is too
well known to need repeating here. Coincident with the more recent
developments in underground distribution certain special situations
have brought about the development of underground cable suitable
for use in high-voltage transmission circuits, inclusive of 132 kv.
Underground installations involving these transmission voltages are
highly special, comparatively few in number and small in extent.
However, they have a definitely favorable effect upon coordination
problems withing the territories surrounding them.
Aerial cable construction for both distribution and transmission
circuits has been used to a limited extent and has a definitely beneficial
effect upon coordination matters. Whether this type of construction
will be extended in the future is not evident.
Grounding of Distribution System Neutrals. — One of the difficult
tasks encountered in distribution systems is that of obtaining adequate
grounding of primary and secondary circuits. Because of this difficulty
the establishment of neutral networks grounded at many points has
become a practise. In most cases in the past, two separate neutral
networks have been provided, one for the primary and one for the
secondary system. However, in several localities these two separate
neutrals have been combined into a common-neutral arrangement pro-
viding in this way an increased multiplicity of ground connections to
both the primary and secondary neutral conductors. Further exten-
sion of the use of this system is probable. This arrangement intro-
duces features of interest from the coordination standpoint, because
of the increased opportunities for the How of currents through the
ground. Experience and investigations so far, however, indicate that
with adequate attention to coordination this arrangement is comparable
in its effect on neighboring communication circuits, to other types of
distribution systems.
180 BELL SYSTEM TECHNICAL JOURNAL
Service Taps on Transmission Lines. — In some rural situations, it has
been found economically impracticable to initiate distribution lines
clue to distances involved. However, in many such situations immedi-
ate electric service is urgently required and in some of these cases,
transmission lines may be located relatively close to the point where
service is desired. In such cases the only alternative to a long distri-
bution line is to tap the high tension transmission line when this can be
done by some less expensive method. Such methods have been devel-
oped and applied to a limited extent. More study and field experience
are needed to determine the effects of these installations on inductive
coordination should they become extensively employed.
Transformer Connections. — In distribution practise, the trend toward
higher primary voltages has been accompanied by the use of the "Y"
connection of the primaries of transformers as a step in the transition
from one voltage class to another. Thus 2300-volt delta systems have
become 2300/4000-volt "Y "connected systems, 6600-volt delta sys-
tems have become 11,000-volt "Y" systems, and the 7620/13,200-
volt "Y" connection is being used. The use of the "Y" connection
of the primary of distribution transformer banks is sometimes necessa-
rily accompanied by a similar connection of the secondary. Such
" YY" connections are usually in urban situations. Also, these banks
usually represent only a small portion of the total transformer capac-
ity on the circuits.
On large transformer banks and in the higher voltages delta-Y con-
nections have long been the prevailing practise. However, where the
" YY" connection is used for purposes of grounding, especial attention
has been given to controlling the effects of this connection in situations
of coordination, and for the absorption of triple harmonic currents it is
common practise to use delta-connected tertiary windings in such in-
stallations. This subject is discussed more fully in another paper in
this symposium.
Wave Shape. — The connection of primary circuits directly to gener-
ating station busses results in service and economic advantages by
eliminating transformations thereby improving voltage regulation and
aiding system simplification. This practise, however, tends to make
coordination more difficult as those harmonics which may be present
in the generated voltage can flow directly out over these circuits.
However, the important bearing of the wave shape of generators and
apparatus of various kinds on the coordination problem has long been
realized and is receiving increasing attention. Even before the forma-
tion of the Joint General Committee the general problem of apparatus
wave shape was being studied both as to the amounts of various har-
TRENDS IN PRACTISE AS AFFECTING COORDINATION 181
monic components which were present in apparatus wave shape and as
to the relative effect of these components when appearing in communi-
cation circuits by induction from power circuits. As a result of this
study an instrument was developed for measuring "Telephone Inter-
ference Factor" of a voltage wave. With this instrument as an aid
a better understanding of the bearing of wave shape has been gained
by the apparatus manufacturers and there has resulted a gradual
improvement in the wave shape of new apparatus.
It is recognized that there is a median line beyond which general
improvement in the inherent wave shape of apparatus would not justify
the attendant increased difficulties of design and increased manufac-
turing costs, — to avoid the alternative of applying in specific cases,
available and less expensive methods of externally correcting wave
shape. Work is now in progress cooperatively between the manufac-
turers and users looking toward the establishing of a measure of wave
shape in apparatus design which will strike an economic balance be-
tween benefits and burdens.
The increasing use of rectifiers for conversion from alternating to
direct current has an influence on inductive coordination. Consider-
able study has been devoted to this matter as result of which methods
for control of the distortion of the d-c. voltage wave caused by the
rectifiers have been applied in several instances and a solution of this
part of the problem appears to be in hand. More study and experience
are needed as regards the specific conditions under which the wave
shape distortion of the alternating current supply would require con-
sideration.
With the progress begin accomplished in the design and application
of apparatus and the better understanding of the influence of circuit
and transformer connections on inductive relations, problems concerned
with wave shape can be expected to steadily decrease. The status
of the cooperative study of this subject is described in this symposium.
Conclusion
A brief outline has been given here of the general trends in plant
development and operating practise in telephone and power systems
with special regard to those trends which affect the problem of coordin-
ation. While naturally there have been influences working favorably
and others working unfavorably toward the problem it is clear that the
preponderant effect of the development now being applied in the two
industries is reducing the proportion of new situations in which specific
coordinative measures are necessary. While to a considerable extent,
as indicated in the body of the paper, this is due to the natural trends
182 HKLL SYSTEM TECHNICAL JOURNAL
of plant design associated with new developments within each of the
industries, it is also true that the extent of the progress made is due in
no small measure to the careful study of all phases of the problem being
conducted by the Joint Committee of the National Electric Light
Association and the Bell Telephone System.
Under the guidance of this Committee and soon after its formation,
the types of situations of physical proximity were classified and certain
broad principles of cooperation were recommended. Soon thereafter
more complete principles and detailed practises were formulated.
These principles and practises were printed and widely distributed tc^
companies and individuals directly interested in the problem of coor-
dination.
The principles and practises thus set up were largely qualitative and
the need for an organized program of research to establish quantitative
data and to develop improved physical facilities for coordination was
early recognized. Accordingly, the Joint Sub-committee on Develop-
ment and Research was organized, and assigned the work of determin-
ing both experimentally and by field experience quantitative data
covering the various aspects of coordination problems, and of devel-
oping detailed methods of effecting physical coordination. Under
this Sub-committee a very large volume of research work has been
undertaken. Results of some of this work have been published and a
considerable amount is now in progress. The three papers to follow
in the symposium discuss much more fully three of the most important
aspects of coordination work at the present time and tell of the work
being done in these fields by the Joint Sub-committee on Development
and Research and by the other branches of the Joint General Commit-
tee's organization.
In reviewing this subject one is impressed by the number of ways in
which the coordination problem touches both the telephone and power
fields, and by the very large amount of cooperative work which has
already been done. This work, as has been indicated, has resulted in
great progress in the satisfactory handling of coordination matters of
all types. This matter concerns two industries both of which are in a
period of rapid development and change, both as regards their size and
as regards the physical arrangements which constitute their plants.
Many new developments in each plant require consideration from the
standpoint of coordination. It is evident, therefore, that if the ground
already gained is to be held and further progress made, the channels of
cooperation between the two industries must be kept in operation
TRENDS JX PRACTISE AS AFFriClTXG COORDTXATIOX l83
both for the consideration of new problems eirising with new develop-
ments in the industries, as well as for the further perfection of the co-
operative methods of handling specific problems. These papers in
other words do not constitute in any sense a final report. They are
intended to show the present status of two very active and rapidly
changing arts and to indicate the highly satisfactory results which have
followed from a number of years of sincere cooperative effort between
the telephone and power industries.
Status of Joint Development and Research on Noise
Frequency Induction *
By H. L. WILLS and O. B. BLACKWELL
The work of finding out the technical facts bearing on the problems of the
physical relations of power and telephone circuits was intrusted to the Joint
Subcommittee on Development and Research of the National Electric
Light Association and the Bell System. This paper has to do with this fact-
finding work so far as it concerns noise frequency induction.
The work on inductive coordination may be classified into three groups of
factors:
L Influence factors which concern the characteristics of the power
circuits.
2. Susceptiveness factors which concern the characteristics of the
communication circuits.
3. Coupling factors which concern the interrelation of power and
communication circuits.
The paper discusses these various factors in detail and describes the work
done by the committee or in progress regarding them. References are given
to published reports and papers which present the results of technical
studies already completed.
Many of the existing noise frequency induction problems have arisen
because of the development of the art of the two industries without such
close cooperation between them as now exists. It is becoming evident, from
the work of this Joint Subcommittee, that while it is not practicable to
design machinery and apparatus for power systems to be entirely free of
harmonics, or to ideally balance either power or telephone circuits, it is
possible to control these factors within limits which, in conjunction with
the control of coupling obtainable by cooperative planning of routes and
coordination of transpositions, permit satisfactory operation of both
services without unduly burdening either.
T
^HE Joint Subcommittee on Development and Research is the
agency through which the National Electric Light Association
and the Bell Telephone System carry out technical work on problems of
physical relations which vitally affect their respective growth and
operating practises. In the present paper and companion papers the
status of this joint development and research work is described.
The present paper, Part II of the Symposium, is concerned with
problems of induction in telephone circuits under normal operating
conditions of power systems which results in noise. Part III of the
Symposium treats of induction at the power system fundamental
frequency, principally that occurring at the time of grounds, short
circuits or other abnormal conditions of power systems. Part IV of
the Symposium treats of the physical relations and of the special noise-
* Part II of the Symposium on Coordination of Power and Telephone Plant.
Presented at the Winter Convention of the A. I. E. E., New York, N. Y., January
26-30, 1931. Published in abridged form in Electrical Engineering, April, 1931.
184
JOINT DEVELOPMENT AND NOISE FREQUENCY INDUCTION 185
frequency and low-frequency problems brought about by the close
proximity of the two types of service when occupying the same poles.
The Joint Subcommittee on Development and Research has sub-
divided its work among ele\en project committees and assigned to
each the actual carrying on of specific research work. Certain of the
project committees are engaged on the problems described in this
paper, while the remainder are concerned with the development and
research problems of the companion papers, Parts III and IV of the
Symposium. The names of these project committees, together with
a statement of the phase of the problem considered by each, is given in
Volume I of "Engineering Reports of the Joint Subcommittee on
Development and Research." ^
Naturally the first steps taken by the Joint Subcommittee were the
review and appraisal of existing information and the exchange of data
between the two interests represented. This paper includes a state-
ment of the problem, with some review of the factors involved, the
results accomplished by the subcommittee and the work projected in
connection with each factor.
Classification of Factors Contributing to Induction
There are certain characteristics of a power circuit with its associated
apparatus that determine the character and intensity of the electric or
magnetic field which is set up in the surrounding medium. These
characteristics are termed "Influence Factors." -
Likewise, there are certain characteristics of a communication cir-
cuit with its associated apparatus which determine its responsiveness
to external electric or magnetic fields. These characteristics are
termed its " Susceptiveness Factors." -
There is a third group of factors which refer to the interrelation of
neighboring power and communication lines by electric or magnetic
induction or both. These are termed "Coupling Factors."'^
Inductive interference is thus the manifestation in the telephone
circuit of a combination of influence, susceptiveness and coupling; and
inductive coordination consists in the control of factors in all three of
these classes to the degree required for satisfactory operation of both
services.
Methods of Control
Physical Separation . — The first method which comes to mind for the
control of inductive eft'ects is that of physical separation obtained by
placing the power and telephone lines on separate routes. A separation
> For references see bibliography.
186 BELL SYSTEM TECHNICAL JOVRXAL
between lines of a few hundred feet practically eliminates the noise-
frecjuency problem whereas the low-frequency problem may exist with
much greater separations. Since the same customers desire both
communication and power services, the two kinds of distribution lines
are necessarily often located on the same streets and highways. Power
transmission lines and toll telephone lines do not, in general, have to be
placed on particular routes and, therefore, separation can often be
employed where such lines are involved. Cooperative advance plan-
ning on the part of the utilities in laying out their plants makes it
possible to employ separation where it is readily feasible and economi-
cal.
Frequency Separation. — Another method of fundamental impor-
tance is the use of frequency separation. By this method, circuits to
be coordinated are arranged so as to be responsive to different fre-
quencies or bands of frequencies, and comparatively unresponsive to
the frequency or band of frequencies employed for the other circuits.
It is thus possible to make many different uses of electricity involving
transmission in the same medium. This solution is familiar to us in
the coordination of radio services.
Fig. 1 shows a diagram of the various uses of the frequency spectrum
for electrical transmission and the manner in which power and com-
munication services are coordinated by means of frequency selectivity.
The first commercial electrical energy available was in the form of
direct current. Shortly thereafter, alternating current was used for
the transmission of power. The nominal frequencies of the current
used for this service in the earlier days range from I673 cycles to 133
cycles. In American practise the frequencies used for power purposes
have practically settled down to either 25 or 60 cycles. There is one
extensive 50-cycle system and a few odd frequency systems. These
latter of 30, ^i, and 40 cycles, and perhaps others, are being rapidly
eliminated, due to the importance of interconnecting them with 60-
cycle systems. At the present time, there is some tendency for the use
of higher frequencies in special machine shop applications. This use,
at present, is principally at 180 cycles and need not concern us here as
its extent is usually confined within a factory building.
In message telephone transmission, the prime consideration is the
transmission of intelligible speech. While the range of response of
the human ear is from about 16 cycles to 15,000 cycles per second,
human speech occupies a narrower range and a still narrower band is
adequate for intelligibility. The present voice-frequency telephone
circuits, especially the longer ones, operate within a frequency band
of about 250 to 2750 cycles per second. The frequency selectivity at
JOINT DEVELOPMENT AND NOISE FREQUENCY INDUCTION 187
the edges of the band is not sharp, however, so that extraneous currents
at frequencies outside of this band may also give rise to noise. This is
POWER SYSTEM USAGE
100.000,000
50,000,000
10,000,000 ^
5,000,000
1,000,000
500,000
POWER
LINE
CARRIER
NORMAL
RANGE
LIMITED
USAGE
10,000
5,000
1,000
500
COMMUNICATION SYSTEM USAGE
FREQUENCY IN
CYCLES PER SECOND
RADIO COMMUNICATION
f RANGE IN WHICH
SHORT-WAVE
TRANSATLANTIC
TELEPHONE
+ CIRCUITS LIE
SHORT
WAVE
BROADCAST
LONG
LONG-WAVE WAVE
TRANSATLANTIC
TELEPHONE
WIRE COMMUNICATION
PRESENT
CARRIER
TELEPHONE
HIGH-SPEED TOOLS —
POWER SUPPLY —
SINGLE POWER SYSTEMS
RAILWAY AND
OTHER POWER SUPPLY
DIRECT-CURRENT
SYSTEMS
lOPEN-WlRE
CARRIER
TELEGRAPH
MESSAGE
TELEPHONE pROGRAM
TELEPHONE
CABLE
CARRIER
TELEGRAPH _ LONG - DISTANCE
RINGING
— TOLL RINGING
— TRAIN CONTROL
SUBSCRIBER
RINGING
DIRECT _
CURRENT _
TELEGRAPH
Fig. 1 — Frequencies used for electrical transmission.
particularly true at the lower end with some of the local exchange cir-
cuits. High quality telephone circuits for program transmission cover
a wider range. This may be on certain circuits as much as from about
188 BELL SYSTEM TECHNICAL JUCRNAL
3>5 to 8000 cycles per second, thus overlapping the fundamental fre-
quencies used for power transmission.
Control of Power Levels. — Coordination by frequency separation
becomes inadequate when the power levels of the various classes of
services differ greatly as with power and telephone services. Thus,
although incidental powers at harmonics of the power circuit funda-
mental frequency are negligible in comparison to the power at the
fundamental frequency, they are large compared to the power em-
ployed in the telephone circuits and fall directly within the frequency
range of the telephone circuits.
While the powers involved in telephone transmission are small as
compared to those on power lines, they are in turn large as compared
to the acoustical power received from the talker or delivered to the
listener. The ordinary telephone transmitter is an amplifier, delivering
to the line several hundred times the voice power which actuates the
diaphram. On the other hand, the receiver requires an electrical
power a hundred or more times that which it delivers as sound to the
listener's ear.
It is obvious that the relative levels of harmonic-frequency power
in the power circuits and voice-frequency power in the telephone cir-
cuits are of major importance in inductive coordination. These con-
siderations have had large influence in the power field in the control
of wave-shape of rotating machinery and transformers, and in the
telephone field in fixing limitations on such factors as wire sizes, spac-
ings of repeaters and instrument efificiencies.
Balance. — Among the most important methods of coordinating
power and communication circuits is the control of their respective
balances to ground and to each other. A power circuit with absolutely
balanced voltages and currents impressed, and with the various con-
ductors arranged in such a way that they would not establish external
electric or magnetic fields, would not have any effect on any type of
neighboring communication line.
Likewise, a telephone circuit in which there were no unbalances and
in which the conductors were arranged in such a way that in the pres-
ence of an electric or magnetic field they would not have any voltages
induced between them would not become noisy from any neighboring
power circuits. Such an ideal state is impossible, but much has been
accomplished by care in the design of the lines and equipment and by
the transpositions of the conductors.
Shielding. — It is possible to materially reduce electric fields by inter-
posing between disturbing and disturbed conductors grounded con-
ductor surfaces known as shields. Magnetic fields can likewise be
JOINT DEVELOPMENT AND NOISE FREQUENCY INDUCTION
189
reduced by interposing conducting paths which circulate current to
set up counter magnetic fields. The power and telephone cables in use
are probably the simplest examples of shielding. A cable sheath is
almost 100 per cent effective as a shield for electric induction, either on
a power cable or on a telephone cable. The sheath is less effective as a
shield for magnetic induction, because of its finite conductivity. It
does not seem feasible at this time to obtain anywhere near perfect
magnetic shielding.
Factors Contributing to Noise Frequency Induction
Influence Factors
There are two characteristics of a power system which are of primary
importance in determining its inductive influence upon neighboring
telephone systems, i.e., its wave-shape and its balance. The wave-
shape is determined by characteristics of apparatus associated with
the system. The balance is determined by the degree of symmetry
of the supply voltages, load impedances, and of the series impedances
and shunt admittances of the lines. While it is not practicable to
design rotating machinery or other apparatus using magnetic cores
entirely free from harmonics, or to realize ideally balanced three-phase
systems, it is practicable to control both these factors within limits
which, in conjunction with a similar degree of control on the coupling
and in the susceptiveness of the communication circuits, permit
satisfactory operation of both services without unduly burdening
either.
The work on influence factors which has been conducted by the
Joint Subcommittee on Development and Research has, therefore, been
directed for the most part toward the study of the wave-shape charac-
teristics of power systems and apparatus and methods for their im-
provement and the investigation of factors affecting the balance of the
power systems and method for their control.
Wave Shape. — In initiating its work on influence factors, the Joint
Subcommittee found little information available as to wave-shape
which might be expected on operating power systems equipped with
various types of apparatus. In order to obtain a broad picture of
wave-shape conditions as they exist in the field, the Subcommittee
conducted an extensive survey of wave-shape conditions on 34 operat-
ing power systems in the eastern half of the country. The program was
arranged to obtain information as to the average and range of magni-
tudes of harmonics present in various types of transmission and distri-
bution systems under normal operating conditions, to observe the
relation between the wave shape of generating machinery under open-
190 BELL SYSTEM TECLINICAL JOURNAL
circuit conditions and under load, to study the efilects of various trans-
former connections on wave shape, and to observe the effects on wave
shape of various types and magnitudes of load.
The measurements made included analyses of the phase-to-neutral
and phase-to-phase voltages and phase currents on a large number of
generators, transmission lines and distribution feeders. Wherever
practicable, data were obtained as to the balance of the operating sys-
tems by measurements of residual voltages and residual currents.
Measurements were also made of the Telephone Interference Factors ^
of the voltages and currents. Where telephone circuit exposures suit-
able for test purposes existed, noise measurements were made on the
communication lines to aid in determining the relation between power-
system wave-shape and balance and telephone circuit noise. The
actual measurements were for the most part conducted by the operat-
ing companies with the cooperation, during the first part of the testing
program, of representatives of the Joint Subcommittee.
The mass of data accumulated during this survey is being summar-
ized in several technical reports which it is anticipated will yield much
valuable information pertaining to the wave-shape problem. An
important practical application of these data will be in connection with
the prediction of wave-shape conditions on new lines which are to be
involved in exposures with communication systems and on which
noise estimates are desired.
In general, the survey data indicate that the magnitudes of the
harmonics present in voltage and current diminish with increasing
frequency, with the exception that a pronounced dip occurs in the re-
gion from 800 to 1500 cycles. This is, no doubt, a result of the efforts
of the machine designers to closely control the harmonics in this im-
portant region. Frequencies above 2000 cycles become extremely
small except where these may be introduced on the power circuits by
superposed carrier communication or signaling services. In general,
the frequencies used for such services have been in the range from 50
to 200 kc, which is above the range employed for carrier communica-
tion on telephone lines.
In the general survey of wave-shape, no efforts were made to select
feeders involved in cases of inductive interference. Aside from the
survey work, however, representatives of the Subcommittee have par-
ticipated in a number of investigations of such cases in which power-
system wave-shape was an important factor. Much valuable data as
to wave-shape conditions under which coordination difficulties are
experienced have been obtained from these studies, while in obtaining
these data the Subcommittee representatives have been of service to
the local companies in the solution of the particular problems.
JOINT DEVELOPMENT AND NOISE FREQUENCY INDUCTION 191
A limited amount of iheoretical work has been carried on having to
do with the effects of load on the harmonics observed in the open-
circuit voltage of rotating machines. This work which was based on
Blondel's two-reaction theory was supplemented by laboratory tests
on several small machines. It was found that the reactions which
take place within the rotating machines, particularly when two or more
are operating in parallel, are so complicated as to practically preclude
accurate computations of the effects. However, the data obtained
from this investigation have been valuable in connection with later
studies.
Balance. — A balanced power circuit is one in which the voltages
between the various phase conductors and ground are equal in magni-
tude and sum up vectorially to zero and in which the phase currents are
also equal in magnitude and sum up vectorially to zero. In a three-
phase system where the currents or voltages are not equal but do sum
up to zero, the currents or voltages can be resolved into two balanced
three-phase systems, one of positive phase sequence and one of negative
phase sequence. In cases where the currents or voltages do not sum
up to zero they contain a single-phase component which is usually
termed residual or zero-phase sequence component. Any three-phase
system can be resolved into its balanced and residual components and
each treated separately. The coupling for the residual components
is usually much larger than for the balanced components and is there-
fore frequently of major importance in coordination problems. Differ-
ences in the magnitudes or departures from phase symmetry of the
three impressed phase-to-neutral voltages, load or line unbalances,
give rise to residual currents or voltages.
Experience has indicated that the outstanding factor in the unbal-
ance of power systems is the existence of triple-harmonic voltages and
currents which may arise either in rotating machinery or in trans-
formers which are connected in star with grounded neutral. Since the
triple-harmonic voltages in the three phase-to-neutral legs are in phase,
they act in a path consisting of the phase conductors and an external
return as, for instance, a metallic neutral or ground.
A large measure of control may be exercised on the magnitudes of
the triple-harmonic residual voltages and currents by the use of certain
transformer connections and by not operating the transformers at high
flux densities.
The magnitudes of triple-harmonic residual currents in grounded-
neutral systems may be minimized by the use of star-delta connected
transformers, in which case nearly all the required triple-harmonic
current circulates in the delta. The opposite extreme occurs with star-
192 BELL SYSTEM TECHNICAL JOURNAL
star connections in which case the full triple-harmonic magnetizing
current flows in the two systems which the transformer interconnects,
the relative magnitudes in each depending on their relative impedances.
Where a star-star bank is connected at one terminal of a line, with a
star-delta at the other, the neutrals at each end being grounded, prac-
tically the entire third harmonic required by the star-star bank may be
expected to circulate in the line connecting the two.
An effective method of control for cases in which star-star connec-
tions are required due to phase relations is the provision of a third set
of windings or tertiaries in the transformers, the impedance of the
tertiaries with respect to the other windings being sufficiently low to
furnish an adequate path for the triple-harmonic magnetizing current.
An alternate method of control, which also provides like phasing on the
two sides of the bank, is the use of zig-zag connected transformers.
In four-wire multi-grounded neutral distribution systems, it has
been found helpful in controlling the residual triple-harmonic currents
from the single-phase load transformers to provide star-delta connected
banks at various points in the network with neutrals connected to the
system neutral. In some cases, these have been three-phase load
banks, in others, special banks installed as a method of control.
The subcommittee is continuing its work on wave shape and balance
through a laboratory study of transformer harmonics and transformer
connections. These tests are being made on small model transformers,
typical of the designs which are used for large sizes on transmission
systems. It is planned to develop the theory applicable to harmonics
from transformers on three-phase systems from the work on these
laboratory models. It is planned to supplement the work by tests
on large transformers in the manufacturer's shops and in the field.
A number of severe noise situations have been created during the
past few years when star-connected generators, operating with
grounded neutral,''-^ have been connected directly or through star-star
transformer banks to transmission or distribution systems. The
interference in these cases resulted from triple-harmonic residual com-
ponents impressed on the system by the particular generator operating
with the grounded neutral. The magnitudes of these currents depend
on the triple-harmonic components in the generator phase-to-neutral
voltage and the impedance to ground of the system. The methods of
control which have been successfully applied in these cases include the
following:
1. Isolating the generator neutral and supplying the system ground
through a suitably designed transformer bank.
2. Grounding the neutral of only those generators designed to be
free from triple harmonics in their phase-to-neutral voltage.
JOINT DEVELOPMENT AND NOISE FREQUENCY INDUCTION 193
3. The use of selective devices such as reactors or anti-resonant
circuits commonly called "wave traps" in the generator neutral for
suppressing the disturbing triple-harmonic components.
Xon-triple harmonic residual voltages and currents may exist from
differences of phase-to-neutral load impedances and from differences
in the capacitances to ground of the three phase wires.
In multi-grounded neutral four-wire systems differences in the single-
phase loads connected between the individual phases and the neutral
may be important sources of residual current. A considerable measure
of control may be exercised by restricting the size of single-phase areas
and balancing the load on the different phases.
Capacitance unbalance to ground may be due to single-phase
branches on three-phase distribution systems. Usually, the more
important effect is that on the single-phase branch where the residual
voltage is practically equal to the phase-to-neutral voltage. The
unbalancing effect on the three-phase system may be minimized by
equalizing the lengths of the branches connected to the several phases.
The residual voltage on the single-phase branch can, where necessary,
be eliminated by the use of isolating transformers or by converting to a
three-phase branch.
Capacitance unbalance may also be due to dissymmetry in the ar-
rangement of the wires of the circuit to each other and to ground.
These unbalances are lowest in triangular configurations of the wires
and largest when all the wires are in the same vertical or horizontal
plane. With multi-circuit lines, a considerable measure of control
may be obtained by suitable phase interconnection of the circuits.
Transpositions are also effective in controlling these unbalances.
CoupUng Factors.— T\iQ coupling between power and communication
circuits is, of course, determined by the degree of their proximity, but
it may be greatly modified by the balance of the two classes of circuits
to each other and with respect to ground. While the most direct and
certain method for reducing coupling is to avoid proximity, means
are available for minimizing the coupling where necessary.
The work on coupling of the Joint Subcommittee on Development
and Research, in the voice and carrier-frequency range, has been
directed toward two objectives: (1) development of improved methods
for predetermining the coupling to be expected in new cases of expo-
sure, and (2) development of improved methods for reducing coupling
for given degrees of proximity.
Several years ago the California Joint Committee on Inductive
Interference ^ completed an extensive series of computations on coeffi-
cients of induction which were expressed in the form of curves for
194 BELL SYSTEM TECHNICAL JOURNAL
various physical relationships of power and telephone lines. These
coefficients indicate the voltages induced in short, isolated, untrans-
posed telephone circuits by unit voltage and current on similarly un-
transposed power circuits. They do not include the small separations
involved with jointly used poles.
These curves and others based on them have been used for many
years in determining relative coupling, when comparing different
exposures, different routes involving various degrees of exposures,
different configurations of power and telephone circuits and for other
comparisons where all factors were substantially equal in the situations
being compared, except those involved in determining the coefficient of
induction. For these purposes they have been very useful. Methods
have not, however, been available whereby these coefficients could be
used for computing noise where transposed circuits were involved and
where many telephone wires were on the line, which exert an important
shielding effect on each other.
The Joint Subcommittee on Development and Research has been
conducting experimental studies both for highway and wider separa-
tions, and those occurring with jointly used poles, so that the effects of
transpositions and of mutual shielding of the many wires involved might
be properly taken into account in determining the noise currents in the
metallic circuits.
In determining the coupling between power and telephone circuits,
it is desirable to differentiate between the effects of the balanced and
residual components of the voltages or currents of the power circuit,
between the effects of voltages and those of currents, and on the tele-
phone line between induced voltage which acts directly in the metallic
circuit, termed "metallic-circuit induction," and that which acts in the
circuit composed of the wires with ground return, termed "longitudinal-
circuit induction."
Since the residual components act in a circuit having ground as one
side with the wires in parallel for the other, while the balanced com-
ponents are confined to the wires of the system, the coupling for the
residual components is much greater than for the balanced components.
The coupling for the balanced components may be reduced by the use
of power-circuit transpositions, while such transpositions have no effect
on coupling for the residual components.
The distance between the power and telephone wires is usually large
as compared to the spacing of the wires of the telephone circuit, so that
the longitudinal induced voltages are large as compared to the metallic-
circuit voltages. The effect of the telephone transpositions being
merely to equalize the relations of the two sides of the telephone circuit
JOINT DEVELOPMENT AND NOISE FREQUENCY INDUCTION 195
to the power circuit, such transpositions do not change the magnitude
of the longitudinal voltages, but do reduce the metallic-circuit voltages.
The relative magnitudes of inducing voltages and currents differ
widely among various power circuits, and may vary greatly with time
on any given circuit. They will also differ considerably at a given
time and on a given circuit among the various frequencies involved.
For this reason it is necessary to consider separately the coupling
arising through the electric and magnetic fields.
X'oltages induced in metallic circuits for the separations between
lines usually encountered are practically proportional to the spacing
of the wires of the telephone circuit. Voltages induced in eight-inch
spaced pairs are thus approximately two-thirds of those induced in
12-inch pairs, while those induced in phantoms on 12-inch spaced side
circuits are twice those induced in the sides. The longitudinal voltages
are, however, practically independent of the wire spacing so that the
contributions which these voltages make to noise in the metallic circuit
are unchanged except as the change in spacing may affect the balance
to ground.
Spacing of the wires on the power circuit and their configuration also
have an important effect on the coupling for the balanced voltages and
currents, the coupling, in general, increasing as the spacing increases.
Coupling for the residual components is, however, affected only to a
minor degree by the spacing and configuration. Much information
bearing on these matters is included in the material on coefficients of
induction published by the California Commission referred to above.
Measurements of coupling have been made by the subcommittee in
a number of situations. These have included cases of (1) exposure of
overhead transmission lines and open-wire toll telephone circuits at
highway separations, (2) overhead distribution lines and subscribers'
telephone cables in joint use and at street separations and (3) overhead
distribution lines and subscribers' open-wire circuits in joint use.
Information was obtained on coupling both for voltages and currents
and for the balanced and residual components. The results of the
work on overhead distribution lines and subscribers' telephone cables
have already been published.^ The other data are to be published as
soon as they are prepared in suitable form.
The work on overhead distribution lines and subscribers' circuits is
relatively complete, covering a wide range of conditions typical of
those encountered in the field. Various arrangements of primary and
secondary conductors covering single-phase and three-phase, three-
wire and three-phase, four-wire systems were investigated. The
shielding effect of the telephone cable was determined and, with the
196 BELL SYSTEM TECHNICAL JOURNAL
open-wire subscribers' telephone circuits, the shielding effect of the
various telephone wires on each other.
For telephone cable circuits when the sheath is grounded at either
one or both ends, the inductive effect of the power circuit voltages on
the wires enclosed is negligible as compared to that of the power circuit
currents. Furthermore, because of the close association of the wires
of a pair in the cable and the frequent twist, the metallic-circuit in-
duced voltages are negligibly small as compared to the longitudinal
\oltages so that, in general, only the magnetic longitudinal coupling
factors are of importance in these situations.
The work further indicates that, for most practical problems involv-
ing overhead distribution lines of the multi-grounded type and sub-
scribers' cable circuits, a knowledge of the coupling for the residual or
unbalanced currents is sufficient, the effect of the balanced currents be-
ing relatively unimportant. However, in cases where the line currents
are particularly heavy or contain exceptionally large harmonic com-
ponents, the balanced currents become important.
In the range of frequencies used for telephone transmission the ratio
of open-circuit voltage induced on a telephone line through electric
induction to inducing voltage on the power circuit is substantially
independent of frequency. When the exposed section of line is con-
nected to the remaining section of the telephone line or to terminal
apparatus, a current is set up which is approximately proportional to
the frequency of the induced voltage. The circuit will perform as if
there were a small condenser connected between the power circuit and
telephone circuit and the induced current experienced will be propor-
tional to the frequency and the magnitude of the inducing voltage on
the power circuit.
The coupling between power and telephone circuits for currents is
in the nature of a mutual inductance, so that the voltage induced in the
telephone circuit is proportional to the magnitude of the inducing cur-
rent in the power circuit and its frequency.
This statement applies strictly only to induction from the balanced
current components. Induction from residual current in the power
circuit is complicated by the effect of the finite conductivity of the
earth. With increasing frequency the earth currents tend to be closer
to the surface and the coupling with the telephone circuit tends to
increase less rapidly than would follow from proportionality with
frequency. The departure from linearity is not large in the frequency
range from 250-2750 for highway separations and for joint use.
Transpositions afford one of the most powerful means available for
controlling coupling of power and open-wire telephone circuits in given
JOINT DEVELOPMENT AND NOISE FREQUENCY INDUCTION 197
situations of proximity. Transpositions operate by neutralizing, in
one section, inductive effects which arise in a closely adjacent section.
It is evident that, in order for transpositions to l)e fully effective, condi-
tions must be substantially alike among the various sections to be
neutralized as regards relations of the power and telephone circuits
to each other, to ground, and among the various circuits on each line.
This latter condition more often applies to the telephone lines, as they
usually comprise many circuits.
These conditions require that balanced and coordinated systems of
transpositions be provided between each point of discontinuity in the
exposure. By "discontinuity" is meant any point at which an impor-
tant change takes place in the physical or electrical conditions of the
circuits, such as loads, branch circuits, series impedances, etc.; any
change in configuration, in the separation of the two classes of circuit
or in their position relative to ground or to some other circuits which
may be associated with either power or telephone circuits closely
enough to appreciably modify the induction.
In addition to meeting these conditions, the telephone transpositions
must also satisfy the requirements for minimizing cross talk among the
various telephone circuits. This, in general, requires telephone trans-
position arrangements of considerable complexity. For this purpose
standard transposition arrangements are available,* adapted for differ-
ent lengths depending upon the distances between the successive dis-
continuities.
In most cases unavoidable irregularities occur in the spacing of poles,
in distances between power and telephone circuits, in presence of
shielding objects, such as trees, and in height of poles, which it is not
possible to treat as discontinuities and take into account in the trans-
position design. In cases where these irregularities are large, the effec-
tiveness of the transposition arrangements is greatly impaired. The
extent to which the effectiveness of such arrangements is imparied due
to these non-uniform conditions is a problem not easily susceptible to
mathematical analysis and reliable information is not now available.
The subcommittee is planning to investigate this problem experi-
mentally by tests on a number of situations involving operating circuits.
Susceptiveness Factors.— The degree to which telephone transmis-
sion is adversely affected by noise-frequency induction depends not only
upon the magnitudes of the induced voltages as determined by influence
and coupling factors, but also upon the susceptiveness factors of the
telephone system. These include the manner in which the induced
voltages and currents are propagated to the circuit terminals together
with the reactions of the circuit unbalances, thus relating the current
198 BELL SYSTEM TECHNICAL JOURNAL
in the terminal apparatus to the induced voltages, the sensitivity of the
receiving apparatus and the operating power level of the telephone
circuits.
Propagation Effects and Balance. — Important differences exist with
respect to propagation effects and balance between open-wire and cable
circuits and between toll and exchange systems.
As pointed out in the discussion of coupling, only the magnetically
induced longitudinal voltages and currents affect telephone cable cir-
cuits. Because of the absence of electric induction and direct metallic-
circuit induction and because of the important shielding effects exerted
by the cable sheath and the various telephone circuits on each other,
telephone cable circuits are much less susceptive than open-wire cir-
cuits.
In open-wire telephone systems consideration must be given both to
electric and magnetic induction and to voltages directly induced in the
metallic circuit as well as to those induced in the longitudinal circuit.
In a line composed of a number of circuits, the currents set up in any
one circuit depend, not only upon the voltage induced in that circuit
and its impedance, but also upon the currents and voltages which are
set up in the rest of the telephone circuits on the line. It is not pos-
sible, therefore, to calculate the induced currents merely from a knowl-
edge of the magnitudes of the currents and voltages on the power
circuits and the coupling between the power circuits and isolated pairs
of wires on the telephone line, considered independently.
These mutual effects among the various telephone circuits exist both
within and without the exposed sections. Thus, the propagation of
the induced voltages and currents along any one circuit is influenced
both by the electrical conditions of this circuit and also by the condi-
tions of all other wires on the line. Additional complexities arise in
the propagation of the induced voltages and currents, because of non-
uniformity in impedances to ground at terminals, points where circuits
join or leave the line, and where lengths of cable may be used at termin-
als or at intermediate points. The impedances to longitudinal induced
voltages and currents vary over a wide range depending on the number
of wires on the line, the relative position of the exposure and the circuit
terminals and the occurrence of sections of cable. Due to reflection
effects from these irregularities, peaks of current and voltage may
exist along the circuits which are large as compared to the correspond-
ing magnitudes at the exposure terminals. If circuit unbalances
happen to exits at these maximum points, metallic-circuit voltages and
currents thereby introduced are increased.
While the distribution of longitudinal voltages and currents among
the various wires upon the telephone line depends uj^on the nature of
JOINT DEVELOPMENT AND NOISE FREQUENCY INDUCTION 199
the inducing field in which it is placed, the experiments of the committee
have shown that a satisfactory degree of approximation for studying
propagation effects can be obtained by energizing all wires on the line
simultaneously at the same potential from a common source. An
extensive experimental study has been made in this way by the
committee in which the magnitudes of the longitudinal voltages and
currents at various points along the line have been measured as well
as metallic-circuit currents set up through the unbalances at the send-
ing and receiving ends of the line.
By making measurements of this sort on a considerable number of
lines of different types of construction and different transposition
arrangements, it is hoped to obtain statistical data whereby the metal-
lic-circuit voltages and currents at the circuit terminals may be deter-
mined from the magnitudes of longitudinal voltages and currents as
measured at exposure terminals.
Unbalances in toll circuits are the result of commercial variation
from the balanced condition, since the circuits are designed to be
symmetrical. These unbalances may consist of resistances in joints,
capacitance or inductance unbalances due to irregularities in trans-
position spacing or to omitted or unspecified transpositions, or differ-
ences in the impedances of apparatus connected in series with the wires
or between them and ground. These unbalances are fortuitous both as
regards their magnitudes and location along the toll circuits. Some in-
crease in importance with frequency and others decrease. These, com-
bined with the irregularities in the propagation of the longitudinal
voltages and currents, cause the resulting metallic-circuit currents in
individual circuits to vary in an erratic fashion with frequency. The
general trend is one of proportionality, independent of frequency within
the important range, between the longitudinal currents and voltages at
the exposure and current in the metallic circuit at the terminals. Tak-
ing into consideration the effects on coupling, the currents at the ter-
minals increase approximately in direct proportion to the frequency of
the inducing voltage or current on the power circuits.
Because of the lower susceptiveness of cable circuits together with
the high degree of balance of the terminal apparatus and because of the
more general use of private rights-of-way, cases of noise-frequency
induction into toll cable circuits have been comparatively infrequent.
For this reason the attention of the subcommittee as far as toll systems
are concerned, has been directed toward open-wire circuits.
In exchange circuits certain inherent unbalances exist due to the
arrangements employed for supervisory signaling, for selective ringing,
and for coin box service. The supervisory system utilizes a low im-
200 BELL SYSTEM TECHNICAL JOl'RNAL
pedance relay connected in series with one side of the central office
interconnecting circuit. The selective ringing scheme involves con-
necting the ringer windings from one side of the line to ground at the
station set. For the coin box service, a coin-collect relay winding is
connected between one side of the station set and ground. These
unbalances have been investigated in detail by the committee and the
results have been published ^ as described later.
The unbalance of party lines due to the ringer ground is usually
much more important than that of the central office interconnecting
circuit due to the supervisory relay. Both are, in general, more impor-
tant than the cable unbalances. Coordination difficulties between
telephone exchange systems and power distribution systems thus
usually involve the party-line circuits before the individual-line circuits
are affected.
The controlling unbalance in the exchange plant when in cable
being in the nature of an inductance between one side of the line and
ground, its importance decreases with increasing frequency of the in-
duced longitudinal voltage. This effect largely counter-balances the
increase in coupling with frequency. Thus, in most situations involv-
ing joint use of poles by distribution circuits and exchange cable tele-
phone circuits, induced currents of the third and fifth harmonics of the
power circuit fundamental frequency assume chief importance.
Exceptions are cases where outstanding harmonics in the range
between 800 and 1500 cycles are present on the power circuits. In
these cases, particularly where the exposures are long, the central office
apparatus unbalances may be more important than those of the party-
line station apparatus.
The method which has been found most generally applicable for
reducing the susceptiveness of exchange cable circuits is the grounding
of the cable sheaths. This reduces through shielding the magnitudes
of the longitudinal voltages and currents. Special station sets having
lower susceptiveness have been used in specific cases where their use
appeared to be the best method.
Power Level and Sensitivity. The magnitudes of the induced currents
in the telephone system having been determined by the influence
factors, the coupling, and the unbalances of the telephone circuits,
the degree to which they impair telephone service depends upon their
intensity as compared to the intensity of the telephone currents.
Consideration has been given by the subcommittee to the possibility
of increases in power levels (a) on local exchange circuits and (b) on toll
circuits. Little promise has been found in the proposal to raise voice
power levels in the local exchange plant as a means of reducing the
JOIST DEVELOPMENT AND NOISE FREQUENCY INDUCTION 201
effects of noise. As previously pointed out, present telephone trans-
mitters materially amplify the jiower received from the voice so that
the electrical power on the telephone line is some hundreds of times
greater than the acoustic power applied. In development work on
telephone transmitters, telephone engineers are proceeding on the
basis that more is to be gained by improving the frequency response of
the transmitter than can be gained by mere increase of power. This
line of development has, of course, the effect of raising power levels at
frequencies where they have been relatively low.
Two proposals for application to the toll telephone plant were
studied. One would involve changing the repeaters now in use at
terminals and at intermediate points to a more powerful type and
equipping all toll circuits with terminal repeaters of this same type.
This would permit raising the power levels without altering the relative
levels of the various telephone circuits and thus would not change the
crosstalk. Another would involve such changes only on certain long
toll circuits, leaving the remainder of the circuits at their present
levels. As the result of a trial installation, it was found that to realize
any appreciable change in level on these circuits, very extensive changes
would be required to avoid crosstalk from the higher level circuits to
the remaining ones which were not changed.
The levels employed in carrier telephone circuits, while somewhat
lower than those used on voice-frequency open-wire telephone circuits
at the receiving end. are higher at the sending ends than the corres-
ponding voice-frequency levels. Since the power system harmonics
in the carrier-frequency range normally are small as compared to those
in the voice-frequency range, carrier-frequency open-wire systems
experience considerably less noise from power systems.
Effects of Noise. — The actual voice power level on telephone circuits
varies over a wide range, depending upon the particular user, his
distance from the telephone central office, and by the transmission loss
in the connection between the two subscribers. Impairment caused
by a given amount of line noise on the circuit may also vary over a
considerable range, depending upon the voice power level and the noise
in the room where the telephone is being used. The method in use
by the Bell System for an engineering basis in considering the effects
of noise on telephone conversation is to substitute for the noise in-
creases in the transmission loss of the circuit. Thus, the circuit with
its actual loss and noise is represented by a circuit of lower noise and
increased transmission loss. These added losses are known as Noise
Transmission Impairments and are abbreviated N. T. I. The N. T. I.'s
were determined from articulation tests and judgment tests made
202 BELL SYSTEM TECHNICAL JOURNAL
on noisy and quiet circuits, and were set up on the basis of typical
talker volumes, transmission equivalents, and amounts of room noise
at the station terminals. Additional transmission loss was added to
the quiet circuits so that noisy and quiet circuits gave equal articulation
or were judged by the observers to be equivalent in their transmission
performance. Thus, the N. T. I.'s are used to indicate an additional
transmission loss or impairment which is occasioned by the presence of
the noise.
The articulation and other tests on which these N. T. I. ratings were
based are now being supplemented by tests conducted under the direc-
tion of the subcommittee. Measurements are being made of the effects
of various magnitudes and sorts of line noise in the presence of typical
amounts of room noise, as determined from a room noise survey made
by the subcommittee, and for representative toll connections and
talker volumes. The line noises being employed are those found typi-
cal from an extensive survey made by the subcommittee on open-wire
toll circuits throughout the country. These tests will afford a basis
for comparing various methods of measuring line noise, including ear
comparison methods now in general use and new visual meter methods
now under development. Thus, this work should lead to a mutually
acceptable method for measuring noise and a basis upon which agree-
ment may be reached as to the impairment in telephone transmission
caused by noise.
Published Results. — As various phases of the technical work being
done are completed, they are published in the form of Engineering
Reports which are released by the Engineering Subcommittee of Na-
tional Electric Light Association and Bell Telephone System. Eight
reports, of which five refer to matters concerning noise-frequency in-
duction, have already been issued. Other reports dealing with this
subject have been recently approved by the Engineering Subcommittee
and will soon be issued. Certain other technical results which have
come from the Subcommittee's work have been presented by various
individuals connected with the work in papers before the A. I. E. E.
Still other results have been published in brief articles in the X. E. L. A.
Bulletin.
One of the problems upon which the technical work of the committee
has been completed and published is that of inductive coordination of
primary distribution systems and exchange telephone circuits in
cable. The results of this work are given in detail in a report ^ entitled
"Minneapolis Joint Use Investigation." This report includes infor-
mation on influence factors applying to various types of power distri-
bution sj'stems, including three-phase, three-wire and three-phase,
JOINT DEVELOPMENT AND NOISE FREQUENCY INDUCTION 203
four-wire systems with various arrangements of neutral grounding,
data on coupling between various typical arrangements of these sys-
tems and telephone cable circuits, and information on susceptiveness
characteristics of telephone systems, including unbalances of lines and
apparatus. To facilitate the use of this information in the day-by-day
coordination problems handled by the operating companies, a summar-
izing report » entitled "Short-Cut Methods for Calculating Noise in
Local Telephone Subscribers' Circuits in Cable Due to Exposures to
Power Distribution Circuits" has been prepared. This report presents
empirical formulas for estimating noise-frequency induction and in-
cludes a brief discussion of the technical factors involved and the
approximations underlying the formulas. Means are described for
reducing influence by the control of triple-harmonic exciting currents
and load unbalances of power distribution circuits, and for reducing
susceptiveness by grounding telephone cable sheaths and by controlling
the unbalances of the telephone station equipment. The information
should be useful to engineers of the operating companies in the coopera-
tive planning of routes to avoid induction troubles.
While a large part of the experimental work connected with the prob-
lem of joint use of local open-wire subscribers' circuits and power dis-
tribution circuits has been completed, the detailed technical reports
have not yet been completed for publication. However, a summar-
izing report ^° which it is believed will largely fill the needs of the engi
neers of operating power and telephone companies has been completed.
This is entitled "Short-Cut Methods for Calculating Noise in Open-
Wire Subscribers' Circuits Due to Joint Use Exposures to Power Dis-
tribution Circuits."
Three reports have been issued dealing with the problem of coordina-
tion of open-wire toll circuits and overhead transmission and distribu-
tion lines. The first ^^ discusses the "Termination of Isolated Ex-
posure Sections to Obtain Normal Metallic-Circuit Currents," which
affords a means of taking into account the shielding effects present
when the line is in normal operating condition. The second report ^- de-
scribes "A Method of Measuring the Balance of Open-Wire Telephone
Circuits with Respect to Longitudinal-Circuit Induction," which should
be useful to the field in the making of special tests and in supplying
statistical data of value for estimating noise effects on open-wire line
circuits. The third report.^^ dealing with "Methods of Measuring
Noise on Open-Wire Toll Circuits," is a detailed presentation of the
various types of tests for studying noise problems on toll lines, and
includes a discussion of the method of analyzing the test data.
Another report '" deals with "The Effects on Inductive Coordination
of Generators Feeding Directly on the Line and Operating with
204 BRLL SYSTF.M TliCIIXICAL JOURNAL
(irouncled Xeutrals." This report includes a detailed discussion of the
factors invoked and describes methods which have been developed
for control of the triple-harmonic residual currents and voltages which
occur with this method of operation.
The results of the work done by the subcommittee on a surxey of
room noise in telephone locations were described in a recent paper.'''
While this was an incidental phase of the general study on effects of
noise on telephone transmission, it was felt to be of timely value,
particularly in respect to the methods of measurement employed.
Using the results of the data obtained in surveys of wave shape on
operating power systems and analyses of noise current on telephone
circuits, a paper ^^ was prepared on the frequency response character-
istics of telephone transmitters and receivers. This paper indicated
that there appeared to be no advantage, in reducing effects of noise,
in shifting the resonance points of telephone transmitters and receivers
from their present region, as the frequency distribution of the noise
currents was such as to give a minimum in this resonance region.
At the time that the joint work was started the need arose for con-
siderable special apparatus to make the measurements which were
required. Some of the important pieces of apparatus for the work in
the voice-frequency range were sensitive single-frequency voltmeters
and ammeters. These needs were taken care of by the development of
sensitive analyzers whereby single-frequency voltages or currents could
be selected from complex wave shapes on either power or telephone
circuits. One form of this apparatus has been described in a paper
before the Institute ^^ and another in a serial report ^^ of the National
Electric Light Association.
In connection with the survey of room noise, a room noise meter was
developed. This was described in the paper ^^ previously referred to
which presented the results of this survey.
Further Work of the Subcommittee
When the subcommittee started its work there was before it an
accumulation of technical problems which had arisen as the arts devel-
oped without such close cooperation as now exists. The statements
given above regarding various phases of the subcommittee's work on
noise-frequency induction indicate the substantial progress which has
been made in the solution of these accumulated problems. They con-
vey also a general picture of the work which the subcommittee has
immediately before it.
It must not be thought, howe\-er, that when these accumulated
problems have been solved the work of the subcommittee will be com-
JOIXT DEVELOPMENT AND NOISE FREQUENCY INDUCTION 205
pleted and its efforts discontinued. This cooperative work must
always bear a relation to the total development efforts of both the
power and communication fields. As has already been pointed out,
this work is concerned with two electrical arts which have been particu-
larly noteworthy for their success in constantly developing their tech-
nical methods and expanding their services. These developments will
surely continue and constant consideration of the physical problems of
coordination is needed to insure that such developments act to steadily
improve rather than to make more difficult the coordination of power
and communication circuits.
Bibliography
1. Engineering Reports of the Joint Subcommittee on Development and Research,
National Electric Light Association and Bell Telephone System, \'ol. 1, 193U.
2. Reports of Joint General Committee of National Electric Light Association and
Bell Telephone System on Physical Relations between Electrical Supply and
Signal Systems, December 9, 1922.
1 "Review of Work of Subcommittee on Wave-Shape Standard of the Standards
Committee," H. S. Osborne, A. L E. E. Trans., \'o1. 38, 1919, p. 261.
4 "Telephone Interference from A-C. Generators Feeding Directly on Line with
Neutral Grounded," J. J. Smith, A. L E. E. Trans., Vol., 49, 1930, p. 798.
5. Engineering Report No. 12, "Engineering Reports of Joint Subcommittee on
Development and Research."
6. Technical Report No. 65, p. 673, book on "Inductive Interference between Elec-
tric Power and Communication Circuits," published by Railroad Commission
of the State of California, April, 1919.
7. Engineering Report No. 6, "Engineering Reports of Joint .Subcommittee on
Development and Research, Vol. I, 1930.
8 "The Design of Transpositions for Parallel Power and Telephone Circuits,"
H. S. Osborne, A. I. E. E. Trans., Vol. 37, 1919, p. 897.
9. Engineering Report No. 9, "Engineering Reports of Joint Subcommittee on
Development and Research."
10. Engineering Report No. 13, "Engineering Reports of Joint Subcommittee on
Development and Research."
11. Engineering Report No. 8, "Engineering Reports of Joint Subcommittee on
Development and Research," \'ol. 1, 1930.
12. Engineering Report No. 10, "Engineering Reports of Joint Subcommittee on
Development and Research."
13. Engineering Report No. 11, "Engineering Reports of Joint Subcommittee on
Development and Research."
14. "A Survey of Room Noise in Telephone Locations," W. J. Williams and R. G.
McCurdy, A. I. E. E. Tr.\ns., Vol. 49, 1930.
15. "The Trend in the Design of Telephone Transmitters and Receivers," N. E. L. A.
Bulletin, August, 1930.
16. "Electrical Wave Analyzers for Power and Telephone .Systems," R. G. McCurdy
and P. W. Blye, A. I. E. E. Trans., 1929, \ol. 48, p. 1167.
17. "Harmonic Analyzer for I'se on Power Circuits," .Serial Report of the Inductive
Coordination Committee, N. E. L. A., January, 1928.
Status of Joint Development and Research on
Low-Frequency Induction *
By R. N. CONWELL and H. S. WARREN
This paper deals with coordination of power and telephone systems with
respect to induction at power system frequency, usually 60 cycles. The
principal problem in this held relates to effects produced under abnormal
conditions on power systems. The factors controlling the magnitude,
frequency of occurrence, duration, and effects, of induced voltages, are
discussed. Different types of protective measures, some applicable to
power systems and others to communication systems, are outlined, including
their respective advantages, limitations, and fields of application. The
reaction on this problem of lightning and of situations involving liabilit}' of
contacts between telephone wires and power wires is touched upon. The
whole matter is treated from the standpoint of the comprehensive joint in-
vestigation of the interference problem which is being conducted by the
N.E.L.A. and the Bell System.
INDUCTION at power system fundamental frequency, commonly
called "low-frequency" induction, has different characteristics and
produces quite different effects from induction at the noise frequencies
discussed in the paper by Messrs. Blackwell and Wills. Smce very
little has been published on low-frequency induction, it seems desirable,
in order to make clear what the Joint Subcommittee on Development
and Research is doing on this subject, to explain the problem in some
detail.
The disturbances in communication circuits due to low-frequency
induction are in general discrete occurrences, coincident with acci-
dental grounds or other faults on neighboring power lines, rather than
being continuous and due to normal power line operation.
Three-phase power circuits, when operating normally, are so nearly
balanced with respect to earth at their fundamental frequency, and
telephone circuits of the ordinary type are relatively so insensitive at
frequencies of 60 or 25 cycles, that induction at these low frequencies
under normal power line conditions is rarely a practical problem. But
when abnormal conditions, particularly faults to ground, occur on
power lines, large unbalanced voltages and currents at fundamental
frequency exist temporarily and at such times there may be induced in
neighboring telephone circuits voltages which are hundreds of times as
great as under normal operating conditions. The induced voltages
under abnormal conditions may reach values sufficient to cause hazard
* Part III of the Symposium on Coordination of Power and Telephone Plant.
Presented at the Winter Convention of the A. I. E. E., New York, X. Y., January
26-30, 1931. Published in abridged form in Eleclrlcal Engineering, April, 1931.
206
JOINT DEVELOPMENT AND LOW-FREQUENCY INDUCTION 207
to telephone employees or interruption to service. Although such ab-
normal conditions occur infrequently and usually last for only the very
short period required to interrupt or clear the power circuit, the effects
which may be produced are so serious that protection against this type
of induction is an outstanding problem in the coordination of power and
telephone systems. A large part of the subcommittee's work has for its
object the development of means for controlling and minimizing such
induced voltages and their effects.
While low-frequency induction is not usually severe except under
abnormal conditions, power circuits which operate at any time on an
unbalanced basis, or which are closely coupled to grounded wires
capable of carrying large currents may, even under normal operating
conditions, create a problem of low-frequency induction in paralleling
telephone circuits in addition to setting up high-frequency disturbances
as explained in the Blackwell-Wills paper. This is particularly true
where the exposed telephone circuits are used for special services such
as the transmission of radio broadcasting programs. Grounded types
of telegraph and other signal circuits also are sensitive to low-frequency
induction.
Classification of Factors Responsible for Inductive
Effects
The same three class of factors which combine to underlie the noise-
frequency problem appear also in the low-frequency problem. As
they appear in the latter, these are:
1. "Influence factors" in the power system, which are concerned
with the magnitude, duration, and frequency of occurrence, of unbal-
anced voltages and currents.
2. "Susceptiveness factors" in the communication system, which
are concerned with the nature and seriousness of the effects produced
by the induced voltages.
3. "Coupling factors" which determine the magnitude of the vol-
tages induced in the communication system, per unit unbalanced
voltage or current of the power system.
In the low-frequency induction problem, the coupling factors are
largely dependent upon the characteristics of the earth and the relations
of power and telephone systems to the earth. If the earth were an
insulator instead of a conductor there could, of course, be no such
thing as fault current in the earth and the coupling between power and
communication circuits would be much less. It would not then be
hazardous for a lineman when in contact with earth to touch a charged
wire. Or if the lines were not in proximity to the earth, there would be
208 BELL SYSTEM TECHNICAL JOURNAL
no chance for a lineman working on the wires to get in contact with
earth. Neither in power systems nor in telephone systems is it actually
necessary that the earth be used as part of an operating circuit but, as
the earth is a conductor, and power and telephone lines and apparatus
are located on its surface, it is essential in both systems that the earth
be taken into account in circuit problems and that paths to earth for
protective purposes be established at certain points.
It must not be assumed, however, that the earth is a perfect conduc-
tor. For the most part the materials of which the earth's crust is
composed are of relatively low conductivity. From numerous meas-
urements in various places the average conductivity over considerable
volumes of earth has been found to range from 10"" to lO"'^ abmho per
cm. cube. The resistance of an earth path is therefore not zero but
may be many ohms or even in some cases many hundreds of ohms.
Most of this resistance is in the immediate vicinity of the electrodes and
can be reduced by increasing the surface area of contact between
electrcde and ground.
In discussing the low-frequency induction problem, it is convenient
to consider the factors controlling:
(1) The magnitude of induced voltages, (2) the frequency of occur-
rence of induced voltages, (3) the duration of induced voltages, and
(4) the effects produced by induced voltages.
Factors Controlling the Magnitude of Induced Voltages
The magnitude of voltages induced in telephone systems in specific
cases depends chiefly on the magnitude of residual currents and voltages
resulting from power circuit faults to ground and on the exposure
conditions.
Residual Currents and Voltages. A balanced power circuit is one
in which the voltages from the various phase conductors to ground are
equal and sum up vectorially to zero and in which the phase currents
also are equal and sum up to zero. Under this condition all the cur-
rents in the circuit are balanced currents and all the voltages are
balanced voltages. If, however, one phase develops a fault to ground,
this relation becomes disturbed, the voltages to ground of the phases
become unequal, and their vector sum, which is the residual voltage
(3 times the so-called uniphase or zero phase sequence voltage) of the
power circuit, is no longer zero. The currents in the three phases like-
wise become unequal and when added vectorially their sum, which is
the residual current (uniphase or zero phase sequence current) of the
power circuit, is no longer zero. In most low-frequency induction
problems residual current is far more important than residual voltage.
JOINT DFA'ELOrMENT AND LOW-FREQUENCY INDUCTION 209
Residual voltages and currents are equivalent to single-phase vol-
tages and currents applied to a circuit consisting of the three line con-
ductors in parallel as one side, and the earth as the other side. Their
large inductive effects are due to the great dimension of the loop formed
by this earth return circuit, much of the return current being effectively
so deep in the earth that its neutralizing action is small. In the case
of the balanced components, the inductive efTect due to the voltage or
current of one conductor is largely neutralized by the voltages or cur-
rents of the other two conductors.
The chief characteristics which determine the magnitude of the resid-
ual voltages and currents are (1) the power circuit voltage, (2) the
impedances of the neutral ground connections, (3) the line and appara-
tus impedances, (4) the fault and earth impedances, (5) the sources of
power supply, (6) the character of ground wires if used, and (7) the
circuit configuration including ground wires.
When a fault occurs between a phase conductor and earth on a power
system having neutral ground connections, these neutral connections,
together with the fault, line conductors and earth, form a closed circuit
for the residual current. Unless the neutral impedance is very high,
e.g., approaching that of an isolated system, the shunting effect of the
capacitance to ground of the line conductors may for most purposes be
neglected and practically the same value of residual current exists at
all points along the line between the fault and the neutral connection
to ground. For simplicity, a system with a single line and single
neutral ground connection may be assumed. With this picture in
mind, it is clear that the value of the neutral impedance may be an
important factor in determining the magnitude of the residual current.
If the fault occurs near the point where the neutral is grounded, the
line and apparatus impedances being low, a small impedance in the
neutral may control the current. On the other hand, for faults occurr-
ing at points remote from where the neutral is grounded, the impedance
in the neutral connection may have to be relatively large to materially
reduce the residual current.
As one limit there is the solidly grounded neutral, i.e., no impedance
is inserted and as good a ground as practicable obtained. This
obviously permits ma.ximum residual current when ground faults
occur. Unless the grounding impedance is very high the residual cur-
rent, and not the residual voltage, is the controlling factor in grounded
neutral systems.
As the other limit there is the isolated neutral, i.e., the impedance
from neutral to earth is infinite. In this case no residual current passes
through the neutral. At the ends of the line the residual current is zero.
210 BELL SYSTEM TECHNICAL JOURNAL
52;raclually increasing to a maximum at the point of fault. The circuit
for residuals is through the capacitance of the line to ground, the mag-
nitude of this capacitance controlling the magnitude of the residual
current, which is much less than with grounded neutral systems except
in cases of double faults when it may be very large. With a single
fault the residual voltage may be a more important factor in respect
to induction than residual current.
The impedance of the fault itself depends upon a number of things,
including the type of line construction and the earth conditions. The
subcommittee has under way investigations to gather data on the
range of fault impedances under different conditions. To determine
the maximum residual current, the fault impedance may be taken as
zero. In many instances, this approximation gives sufficiently close
results, particularly if the fault is remote from the grounded neutral
so that line, neutral and apparatus impedances are controlling. In
case of conductors falling upon the ground, local earth conditions
largely determine the fault impedance. On a steel tower line an insula-
tor breakdown results in a relatively short arcing path to grounded
metal, whereas, in wood pole construction, the pole itself introduces
considerable impedance unless nullified by guys or other metal.
The foregoing discussion of residual current has been confined prac-
tically to the situation brought about by single faults to ground.
Double faults at separate locations sometimes occur and these are
equivalent to a phase-to-phase short circuit through the earth, giving a
large residual current in the intervening section of line. If the two
faults in such a case are on opposite sides of an exposure, very severe
induction may result. Experience shows that double faults at separate
locations constitute only a few per cent of the total faults occurring on
grounded-neutral power systems but are a much larger percentage of
the total faults on systems normally isolated from ground.
The presence of ground wires on a line may have considerable in-
fluence on fault impedance. Being connected to ground at frequent
intervals, such wires decrease the impedance to ground where a break-
down occurs between a phase conductor and a ground wire or any
metal in contact with a ground wire. A ground wire tends to increase
the total residual current but on the other hand its controlling function,
from the induction standpoint, is that of a shielding conductor tending
to decrease the induced voltage.
Circuit configuration does not have a large influence on unbalances
due to abnormal conditions, but it has an important eftect upon any
unbalance of a power circuit under normal operating conditions. To
be balanced, the phases of the power circuit must be symmetrical with
JOINT DEVELOPMENT AND LOW- FREQUENCY INDUCTION 211
respect to each other and to earth. To the extent that the capacitances
and inductances of the several phase conductors differ, residual voltages
and currents will result. Transpositions afford a means for compen-
sating for these circuit unbalances.
In cases for which protective measures are being considered, it is
important to be able to estimate the magnitude of the residual current
when faults occur at different points on the power system. Apart from
inductive effects, this is a question of importance to power companies,
since forecast of currents under different fault conditions is essential
in the design and setting of protective relays. Much work has there-
fore been done by different investigators on methods of predetermining
these currents. Helpful mathematical methods have been developed,
though sometimes the results obtained by their use are open to question
due to lack of accurate values of some of the important impedances.
Proper allowance for fault impedance and the effect of ground wires is
sometimes difficult to determine and in cases of complicated networks
approximations usually have to be made. To facilitate the numerical
computations, calculating boards of varying degrees of elaborateness
have been developed. The subcommittee is investigating this matter
and by experimental work is checking the results of estimates and
acquiring further knowledge of the range of the variable factors.
Through this work, it is hoped to increase the convenience and accuracy
of these important computations.
Exposure Conditions.— The relationship between power and tele-
phone lines with respect to the exposure conditions is defined by the
"coupling coefficient" or "coeffiicient of induction," a factor which,
when multiplied by the value of current (or voltage) in the power line,
gives the resulting voltage set up in the telephone line. A power line
and a neighboring telephone line have several different coupling
coefficients corresponding to different conditions, such as, whether
the induced voltages are due to power current or power voltage, to
balanced or residual components, and whether they are voltages in-
duced along the conductors (or to ground) or are induced directly in the
metallic circuit. Low-frequency induction is predominantly magnetic
in character and the coupling which is most significant is that between
the power conductors and the telephone conductors, both considered
with earth return. The induced voltages are due principally to
"longitudinal circuit induction."
A number of dimensional factors affect the magnitude of this
coupling, such as the length of the exposure, the separation between
lines, and the locations of ground connections on the two systems.
Local conditions as to earth conductivity and the arrangement of
212 BELL SYSTEM TECHNICAL JOURNAL
geological strata for some distance below the earth's surface, constitute
other important factors. An accurate mathematical evaluation of the
coupling between earth return circuits is difficult. Formulas have been
developed under simplifying assumptions as to symmetry and homo-
geneity, which aid in explaining and interpreting experimental results
and in predicting approximate values of coupling in cases where experi-
mental measurements are not available.
Assuming uniformity of exposure conditions the coupling varies
directly with length of parallelism, except for end effects or interactions
between ground connections of the two lines. Increase in separation of
the lines diminishes the coupling but the exact relationship depends
upon the distribution of current in the earth which in turn depends
upon the frequency and the earth conductivity. In many cases differ-
ent strata of different conductivities are involved in the path of the
earth current, which adds to the difficulty of correlating experimental
and theoretical results. The effect of earth conductivity on coupling
is accentuated as the lines are more widely separated. At roadway
separation, large differences in earth conductivity affect the coupling
only moderately; but at separations of one half mile to one mile,
coupling values may differ by 20 to 1 or more, due to the range in
v^alue of earth conductivity. Irregularities in exposure conditions
such as changes in direction of one or both lines, crossovers, and angular
exposures of varying separation, are complications which frequently
occur in practise.
The voltages set up in neighboring communication circuits by power
currents are due usually to inductive coupling but in some cases are
due partly or wholly to resistive coupling. It is seldom necessary in
practical studies to try to segregate these two components of voltage,
since their effects in the telephone system are not a function of the
phase relationships of these components to the power line current which
produces them. It is not unusual to speak of inductive coupling as
including both inductive and resistive coupling.
Any grounded circuit in proximity to power and telephone lines
within an exposure brings about a certain amount of shielding through
the reaction of the currents induced in this conductor upon the primary
magnetic field set up by the residual current in the power circuit. In
this respect a shield wire acts like a short-circuited turn on a trans-
former. The effectiveness of the shielding depends upon the conduc-
tance of the shield wire, the manner and effectiveness of its grounding,
and its position with respect to the power and telephone wires. Such
a wire affords maximum shielding when closely coupled to either the
power wires or the telejihone wires, when its conductance is high, and
when its ground connections are of low resistance.
JOINT DEVELOPMENT AND LOW-FREQUENCY INDUCTION 213
The variation of coupling with separation and with earth conditions
is of great practical importance in the coordinated location of lines.
Most of the subcommittee's study of coupling, therefore, involves
field investigations of the variation of coupling with separation under
different earth conditions, and is furthermore directed toward devising
convenient and accurate methods of predetermining coupling in practi-
cal cases. Also, by studying and correlating experimental data derived
under different conditions and from widely separated parts of the coun-
try, the subcommittee hopes to arrive at a better empirical basis for
estimating coupling. Some of the work on this subject has already
been presented.^
Factors Controlling the Frequency of Occurrence of
Induced Voltages
The frequency of occurrence of induced voltages in paralleling com-
munication lines, while chiefly dependent on the frequency of occur-
rence of faults on the power line, is also somewhat affected by the loca-
tion of the exposure with respect to the location of neutral grounding
points. For example, if there is only one neutral ground, faults occur-
ring between it and the exposure will produce relatively little induced
voltage.
The frequency with which faults occur is usually traceable to features
of electrical and mechanical design, the character and amount of in-
sulation, and the location of the lines. Specifically, the factors which
appear to be responsible for the majority of faults on power lines are:
poor configuration, inadequate spacing and clearances, inferior insula-
tion, lightning, fog, smoke and dirt, birds and animals, proximity of
lines to external objects apt to interfere with operation mechanically
or electrically, and certain mechanical features of design affecting the
strength of construction, such as ineffective anchors, guys, or conduc-
tor and ground wire supports, particularly at angles and dead-ends, and
insufiicient bearing areas of subsurface structures.
Factors Controlling the Duration of Induced Voltages
The length of time faults are permitted to remain on a power system
is controlled by the kind of protective relaying employed and by the
type and condition of the circuit breakers and other terminal equip-
ment. The type of relay system, the degree of sectionalization ob-
tained, the adequacy of the circuit breaker as to speed and rupturing
capacity, and the maintenance of the equipment are the most impor-
tant factors.
' For references see bibliography.
214 BELL SYSTEM TECHNICAL JOURNAL
Generally, the type of fault has little effect on the duration if there is
sufificient current to operate the relays. Conditions have been noted,
however, where the fault is of such high impedance that the current is
not adequate for the operation of the relays. Such high impedance
faults usually occur on wood pole lines and may result in burning of
pins, crossarms and poles. They may also occur on steel tower lines
as the result of branches of trees getting in contact with conductors.
Effects Produced by Induced Voltages
Low-frequency and transient voltages induced on telephone circuits
m^iy produce a variety of effects depending upon their magnitude and
duration. These effects include service interruption, false signals, tele-
graph signal distortion, damage to plant, electric shock, and acoustic
shock.
Telephone circuits are very low energy circuits, the voltage for talk-
ing purposes rarely exceeding one or two volts, with maximum current
measured in milliamperes. For signaling purposes a maximum of 165
volts peak, is used with currents limited to about 0.10 ampere. For
telegraph service the voltages are limited to 135 volts between wire and
ground, while the current is limited to less than 0.10 ampere. By
contrast, the voltages due to induction, in some cases of exposure, may
be a thousand volts or more.
Service Interriiption. — When the telephone protectors are operated
by induced voltage the behavior of the protector discharge gaps de-
pends upon the magnitude of the voltage and current and the length
of time the discharge lasts. In cases where the discharge is not
promptly extinguished or where the current is very high, the discharge
gaps may become permanently grounded. This causes interruption
to service until the affected protectors can be replaced, the time neces-
sary for such replacement depending, of course, upon the protector
locations.
False Signals. — False switchboard signals are likely to be coincident
with protector operation. They produce a bad service reaction due
to operators answering false calling signals and cutting off connections
because of false disconnect indications.
Distortion of Telegraph Signals. — The induced voltages appear in
just the same paths over the wires as the operating voltages of grounded
telegraph. The effect of such induced voltages depends on their
magnitude, character, and duration. Voltages much lower than those
sufficient to operate the protectors may cause detrimental effects
ranging from a slowing down of speed to complete failure. Where the
duration is short, the effect may be limited to distortion of signals, or,
if the voltages are high enough, to momentary interruptions.
JOINT DE VELOPMENT A ND LO W-FREQ UENC Y IND UCTION 2 1 5
Damage to Central Office or Other Telephone Plant. — The dielectric
strength of the telephone plant is adequate for the voltages used in
communication service, with appropriate factors of safety, but higher
voltages may sometimes, notwithstanding the protective devices, cause
dielectric failure, thus damaging the plant, particularly cables and wir-
ing or apparatus in telephone offices.
Electric Shock. — Telephone linemen in the course of their work
upon wires at relatively close spacing, cannot avoid getting in contact
with the wires and if the wires were subject to sufficient induced voltage,
the men would be liable to receive electric shocks. On severely ex-
posed lines such voltages are liable to occur at any time, suddenly and
without warning. Electric shock might either inj ure a lineman directly
or startle him and cause him to lose his hold and fall from the pole.
Voltage to ground due to induction appears not only within the ex-
posed section of line but considerably beyond. A similar, and in some
respects worse, condition may exist with respect to employees working
on cable circuits which are either exposed or directly connected to
exposed circuits. In cables the wires on which the foreign voltage
appears are very close to the grounded metal sheath and usually also to
other wires at approximately earth potential, as well as to the earth
itself. This problem has become more difficult with the rapid growth
of the telephone and electric power systems and is engaging the sub-
committee's serious attention.
Acoustic Shock. — Acoustic shocks are liable to occur with the break-
down of telephone protector discharge gaps, which temporarily un-
balances the circuit and causes a sudden and abnormally large current
in the receivers. This current gives rise to sudden and severe flexures
of the receiver diaphram, which produce loud sharp noises in the ear of
a person using the receiver. Telephone operators, due to the nature
of their work, are particularly liable to acoustic shocks, the effects of
which range from minor reactions to severe general disturbances of the
nervous system which may be painful and of long duration. In
addition, if danger of severe shocks exists, the operating force may be-
come fearful and the impaired morale seriously affect the service.
Types of Protective Measures
The foregoing effects of induction from paralleling power lines may
be reduced by: (1) measures in the power system to limit the influence,
(2) measures in the communication system to limit the susceptiveness
and (3) coordinated location of lines or other means to reduce the coup-
ling. As a solution in a specific situation, one measure may be suffi-
cient or two or more measures may be required, depending on the con-
216 BELL SYSTEM TECHNICAL JOURNAL
ditions. The solution should afford the necessary protection without
hampering the development or operation of either system. Where
there are two or more alternative solutions, the one which is best from
the engineering standpoint, including both the technical and economic
aspects, should of course be applied.
Cooperative planning in advance of construction is especially impor-
tant in situations involving low-frequency induction, because of the
wide ranges in magnitude both of coupling factors and of residual
currents. By advance notifications of construction it is possible to
bring up for analysis the low-frequency effects which the proposed
construction would bring about and, if necessary, to agree upon changes
in the plans to prevent or reduce these effects.
As to the physical dimensions and relations of power and telephone
lines which constitute an exposure there are no blanket rules for guid-
ance; each case requires specific consideration. Due to differences in
geological conditions and other variable factors, a given length of
parallelism at a given separation might give satisfactory results in one
location, whereas an exactly similar physical relationship of lines in
another location might result in the communication system being
rendered inoperative at times of power system fault. This fact
emphasizes the necessity of advance planning and cooperative study of
situations as they arise. Such cooperation may easily lead to a satis-
actory solution of situations which at first seem very difficult. On the
other hand situations which at first appear devoid of any possibilities of
trouble may on careful study be found to require protective measures.
Protective Measures for Poiver Systems. — It will be evident from the
foregoing discussion that protective measures to reduce the inductive
influence of power systems should be directed to limiting the magni-
tudes of unbalanced currents and voltages, particularly under abnor-
mal conditions, and to reducing the duration and frequency of occur-
rence of abnormal conditions. Of such protective measures some are
concerned with fundamental questions of line and system design and
must be incorporated in the construction plans, while other measures
are of such a character that they may either be incorporated in the
original construction or added later if found necessary as a result of
subsequent experience or developments in either the power or tele-
phone system.
Fault-Resistive Design and Construction. — As mentioned in the paper
by Messrs. Harrison and Silver the methods employed in reducing
the frequency of occurrence of faults are primarily involved in the
design and construction of the power line, i.e., adequate insulation,
clearances, and spacings, and so arranging the component parts of the
JOIXT DE VELOPMENT A ND LOW-FREQ UENC V IND UCTIOX 2 1 7
structure that the Hue will in effect be fault- resistive. Increasing
demands for better service by the public combine with considerations
of inductive coordination to justify greater attention to fault-resistive
line construction.
F"aults may result from improper guying of poles, i.e., guys so located
that the spacing between guys and conductors is inadequate, or the
path from insulator to crossarm brace and thence to the guy is insuffi-
cient to withstand the voltages imposed. The conductor spacing may
be inadequate or the configuration of the circuits may be such that the
sudden unloading of conductors coated with sleet will result in their
whipping together, or, if a ground wire is used, it may be so located that
the unloading of sleet will cause the conductors to whip into the ground
wire, or the design of the line, either steel tower or wood pole, may be
such that inadequate strength is provided for the mechanical loads
incurred.
Attention is being given to the location of lines as a material factor
in limiting the number of outages resulting from external sources, such
as lightning, broken trees, blasting, and automobiles. For example,
lines built in valleys are less subject to failures due to lightning and
wind storms than lines built over hills.
There is little need to call attention to the grade of insulation em-
ployed on power lines as recent lightning studies and papers have
emphasized the importance of rationalization of insulation throughout
the plant. By this method it is hoped that preferential points of
failure would be established, thus permitting prompt restoration of
service without damage to expensive equipment since most of the faults
would be confined to the line.
The amount of insulation to be employed on lines is aftected by
topographical and climatic conditions. Lines in areas relatively free
from lightning or shielded from lightning disturbances may, of course,
employ less insulation without increasing the number of faults. On
the other hand, lines built in areas where lightning is prevalent may
justify not only higher insulation but also, on steel tower lines, the use
of ground wires as an additional protection. Areas where salt fog,
smoke, or chemical fumes are prevalent require special treatment as to
the form of insulation used.
Laboratory tests and limited field experience indicate that a proper
utilization of the inherent insulating properties of wood in structures
may result in considerable improvement in line operation. The sub-
committee is investigating the service performance of wood pole lines
of differing designs with a view to determining how much may be ac-
complished in reducing the number and severity of faults by suitable
218 BELL SYSTEM TECHNICAL JOURNAL
arrangements of metal braces, fittings, guys, etc., to avoid so far as
possible shunting out the insulation of the wood.
To the experienced designer the protective measures to be employed
on lines subject to frequent faults are obvious, namely, the rearrange-
ment and reconstruction of the tower or pole top to obtain greater
spacing between conductors or greater clearance between conductors
and other metal parts. In some cases spacing and clearances would be
materially improved by utilizing a triangular configuration so that
the conductors are not likely to come in contact with each other or the
ground wire when sleet or other conditions cause whipping or dancing
of the conductors. In other cases, merely a relocation of the point of
attachment of guys would improve conditions without materially
decreasing the strength of the structure.
Fault-Current Limiting Measures. — Resistors, or reactors, in the
neutral ground connection of a power system provide a means of
directly limiting the magnitude of the residual currents, except in cases
of double faults. In cases where the residual currents can be so far
reduced as not to set up induced voltages of high values in the com-
munication system without reacting unfavorably on power system
operation, this method alone may afford a satisfactory solution. In
such cases it has the further advantage of reducing the stresses to the
power system due to the fault current. Where it is impracticable to
clear up a situation by residual current limitation alone, this method
may be effectively used in combination with other protective measures.
The reduction in residual current which will be brought about by
adding a given amount of impedance in a neutral ground connection
can be estimated with reasonable precision. It is not so much this
question therefore, that requires study by the subcommittee as it is
the question of the limitations and costs of this protective measure,
and its reaction upon the power system. Included in this work is a
study of the relative advantages of inductance as compared with resist-
ance for accomplishing such current limitation. The subcommittee
has under observation a number of installations of current-limiting
devices and is engaged in experimental and theoretical studies and in
field observations by means of recording instruments to determine the
possibilities of this type of protection.
In non-grounded power systems a single fault on a phase conductor
results in the charging current of the system flowing to earth through
the fault. The other phases, rising to full line voltage above the
grounded phase, create a system unbalance which may manifest itself
by induction in paralleling communication lines. In such cases the
problem is one of electric induction except for the magnetic induction
set up by the charging current.
JOINT DEVELOPMENT AND LOW-FREQUENCY INDUCTION 219
When double faults occur on either grounded or nongrounded sys-
tems, severe magnetic induction is liable to result and under these
conditions it is difficult to limit the residual current.
Shielding. Ground wires on a power line, while tending to increase
the total residual current, serve the purpose of shielding by reducing
the strength of the external electric and magnetic fields set up by the
residual voltages and currents. The net effect of ground wires from
the low-frequency standpoint is to reduce the voltages induced in
paralleling communication circuits under abnormal power circuit con-
ditions. The effectiveness of such shielding depends on the impedance
of the shielding conductor and its ground connections. Under favor-
able conditions the induced voltage at 60 cycles in paralleling commun-
ication circuits may be reduced about 40 per cent by this method.
Such ground wires, if used on wood pole lines, have a disadvantage in
that they impair to some extent the insulating property of the poles.
High-Speed Circuit Breakers and Relays. —Very sensitive high-speed
relay systems have been developed which, together with high-speed
types of circuit breakers reduce the time duration of a power line fault
to approximately 1/10 second, as compared with one half second to
three seconds required by the older forms of relays and circuit breakers,
thus tending to minimize the effects of induction. On the other hand
inadequate relaying, or the omission of automatic circuit breakers,
may extend the duration of faults to a point where the hazards to
power apparatus are serious. High-speed breakers and relays are
expensive and it is difficult to justify them solely as a remedial measure
for induction, particularly as the speeds of operation now available
for relays and breakers on power systems, have not reached values
which make them a complete solution of coordination problems.
However, with the increasing size and interconnection of power sys-
tems, high-speed relays and circuit breakers are playing and increas-
ingly important part in promoting power system stability.
Periodic testing of relays and circuit breakers accompanied by com-
plete overhauling at regular intervals, will do much to reduce the dura-
tion of faults and to prevent improper functioning of the equipment.
The subcommittee is following the developments in high-speed
breakers and relays with much interest. If such devices should come
into general use for all classes of service it is expected that they would
materially improve the whole inductive situation.
Improvement in Balance. — As mentioned above, low-frequency in-
duction between power and communication lines is sometimes exper-
ienced under normal operating conditions. On grounded telegraph and
signal lines the trouble usually manifests itself by a chattering of tele-
220 BELL SYSTEM TECHNICAL JOURNAL
graph instruments or by false signals. Improvement in balance of the
power line by transpositions will in some cases correct the difficulty.
Protective Measures for Communication Systems. — In general, meas-
ures applicable to the communication system to prevent or reduce the
effects of induced voltages take the form of arrangements or devices
for removing or counteracting the voltages to ground or the currents in
the telephone circuits which might be produced by the induced voltages.
Bell System Standard Protectors. — It is Bell System standard practise
to equip all telephone circuits which are exposed to the liability of
foreign voltages, with electrical protective devices. These devices are
made in various forms and combinations for different plant and ex-
posure conditions. The protector used at central offices and at sub-
scribers' stations includes a discharge gap which operates at approxi-
mately 350 volts and a fuse which opens the circuit at about 10 amperes.
Such devices are intended to offer a measure of protection against
lightning discharges and against the voltages and currents resulting
from accidental contacts with foreign wires or from low-frequency in-
duction.
In order to protect telephone linemen or others working on open-
wire lines against electric shock from induced voltages, it is necessary
that the voltages between line wires, and between each line wire and
ground, be kept low. The use of protectors at central offices does not
so protect the linemen as the impedance drop on the line wires permits
high voltages between wires and ground at other points, such as the
terminals of the exposed section.
It appeared however, that protectors of the Bell standard type
might be used on open- wire lines at locations immediately adjacent
to exposures to limit induced voltages to ground. A number of in-
stallations of this kind have been made but observations over a
period of time show that they introduce serious troubles as the pro-
tectors, being subjected to heavy discharges, often become permanently
grounded thus interrupting service. It also sometimes happens, as all
the line wires are not always equally exposed, that some of the protec-
tors operate and others do not, resulting in objectionable voltages
between line wires.
Relay Protectors.- — In view of the inadequacy of existing forms of
protectors for such use, the subcommittee is experimenting with a
"relay protector." This device includes Bell standard protectors in
combination with a relay which operates to short-circuit them upon
the occurrence of a discharge, thus relieving the protectors of the duty
of carrying the large discharge current and greatly reducing their
tendency to become permanently grounded. In more recent types all
JOINT DE VELOPMENT A ND LOW-FREQ UENC Y IND UCTIO N 22 1
the relays at a protector point are electrically interlocked, so that
when any relay operates all line wires are grounded within a few cycles.
Several trial installations of relay protectors have been made and
are under observation. To guard against voltages to ground within
the exposure these protectors have to be placed within, as well as at
the ends of, the exposed section of line. Where the longitudinal
induced voltage is large, protectors are required at a number of points
within the exposed section.
The effective application of such protectors requires grounds of the
order of one or two ohms and an important feature of the investigation
is to devise methods of constructing and maintaining such grounds at
remote points along the line.
The subcommittee is investigating in the field and in the laboratory
the effectiveness, cost, reaction on service, and other practical ques-
tions relating to the installation and maintenance of this method of
protection.
Acoustic Shock Reducers. — Since acoustic shock due to induced vol-
tages involves dissymmetrical discharges across the two sides of the
protector, efforts have been made to devise a protector which would
break down and discharge symmetrically, i.e., provide two reliable
low-impedance paths for heavy discharges, which would at all times
have very closely the same arcing impedance. Thus far the subcom-
mittee has not been successful in developing a practicable protector of
this kind.
For the purpose of equalizing the voltages on the protector during
the discharge period, an accessory device termed a "discharge balance
coil" is under investigation. It consists of two equal windings on a
common core, each in series with the discharge gap of one side of the
line, and so arranged that the fluxes set up by the circuits in the two
windings are in opposition. The "booster" action of this coil tends to
equalize the discharge currents. This reduces acoustic shock from
induced voltages, provided all protectors are so equipped and the line
itself has no large unbalances. When however, voltage is impressed
on one wire only of a telephone circuit, as by accidental contact, these
coils have a detrimental effect on the action of the protector in reducing
voltage to ground, as they introduce impedance in the protector dis-
charge path.
Development work is also being conducted on other types of acoustic
shock reducing measures which do not attempt to prevent unbalanced
current but merely to shunt it out of the telephone receiving circuit.
Obviously a device acting on this principle to be successful must be
practically instantaneous in operation. One of the most promising of
222 BELL SYSTEM TECHNICAL JOURNAL
such devices consists of a high ratio step-up transformer with its pri-
mary connected directly across the receiver to be protected. The
secondary is connected to a low voltage discharge gap. Any abnormal
voltage across the primary operates the discharge gap and the trans-
former becomes a low-impedance shunt. A number of field trials of
these reducers applied to operator's receivers have been made. While
not affording the full degree of protection desired they have been found
to reduce substantially the severity of acoustic shocks and it is believed
that they will be of considerable benefit in cases where some form of
protection against acoustic shocks to operators is urgently required.
Another device based on the shunting principle consists of opposingly
poled copper oxide rectifiers connected across the receiver. These have
the property of greatly diminishing impedance with increasing voltage.
The problem is to obtain a sufficiently sharp change in impedance
with voltage, while avoiding a normal impedance so low as to cause
serious transmission losses. As an aid to this end, biasing batteries are
under investigation.
The committee has also investigated the saturating characteristics of
a vacuum tube for acoustic shock reduction. The properties of a
vacuum tube are such that the output current cannot be increased sub-
stantially beyond a definite value regardless of the input voltage. This
feature can be made use of to limit shocks by a design which will pass
currents substantially without distortion up to approximately the
highest value of signal current used, thus cutting down the shock vol-
tages which exceed the normal signals. While quite effective, this
method involves apparatus which is more bulky and expensive than
the transformer and spark-gap type reducer. Telephone repeaters
accomplish this result to some extent and are being investigated by the
subcommittee, to determine the quantitative reduction of acoustic
shock by this means under practical conditions.
In cases where toll or trunk lines are exposed, an acoustic shock
reducing device which could be placed at the ends of the lines would
have the advantage of protecting subscribers as well as operators.
Development work to obviate certain difficulties in using such a device
is under way.
An effort is being made to develop a telephone receiver which will
saturate between the values of current required for effective speech
transmission and values of current which produce acoustic shock.
This requires a sharp bend in the saturation curve of the iron employed
in the receiver magnetic circuit. Until the development of permalloy,
this feature was not approachable, but experimental permalloy re-
ceivers have now been developed, and, while it has not yet been possible
JOINT DEVELOPMENT AND LOW-FREQUENCY INDUCTION 223
to achieve the end sought without serious sacrifice in transmission,
work along this line is continuing.
Improved Insidation. — A slight reduction of susceptiveness to inter-
ference by low-frequency induction could be secured by providing
increased dielectric strength to ground in communication circuits and
their associated apparatus. Another method would be to insulate or
isolate all conducting parts of the communication system so as to pre-
vent contact by employees or others with wires or apparatus which
may carry a dangerous voltage. Neither of these appear practicable
at this time.
Drainage. — Drainage is a method for controlling the parts of the
circuit in which the induced voltages appear and causing these voltages
to be consumed in those parts where they are least harmful. This is
accomplished by connecting the telephone conductors to ground,
preferably through balanced impedance coils, at certain points through-
out the exposure. Assuming low resistance grounds at the drainage
points, the resulting voltage to ground at such a point after drainage
is established is limited to a value corresponding to the voltage drop
over the impedance of the coil and ground connection. If this im-
pedance is small compared to the other impedances in the drainage
section, the voltage to ground at the drainage point is a small part of
the total voltage induced in that section.
Under present conditions, the application of drainage is limited to
special situations where interference with circuit testing and main-
tenance is of relatively minor importance and where superposed d-c.
telegraph and carrier telephone are not used.
Neutralizing Transformers. — The neutralizing transformer is a device
for introducing into an exposed communication wire a voltage in op-
position to the voltage induced by the disturbing circuit, thereby to a
certain extent neutralizing the latter. The neutralization is effected by
means of transformer action, the primary coils of the neutralizing
transformer being connected to conductors which are grounded at the
terminals of the exposure (or section of exposure), so that the voltage
induced in these conductors will send currents through the transformer
primaries. These primary currents induce in the secondaries of the
transformers voltages substantially in opposite phase to the voltages
induced in the telephone wires by the power circuit. The secondaries
being connected in series with the exposed communication wires, the
neutralizing action is obtained.
On account of introducing crosstalk and adversely affecting tele-
phone transmission and carrier, application of neutralizing transformers
has been confined chiefly to telegraph circuits. No applications of
224 BELL SYSTEM TECHNICAL JOURNAL
these devices to power line exposures have been made. They are, how-
ever, being studied by the subcommittee to see whether the objections
mentioned above can be overcome and to determine their possible
field of application.
Shielding. — Shielding on a telephone line may be effected by special
grounded conductors, by working conductors, or by cable sheaths.
Miscellaneous structures such as pipe lines or rails in the immediate
vicinity of an exposure also introduce more or less shielding. The
employment on a telephone line of a high conductance shield wire,
well grounded at the ends of the exposure and at intermediate points,
may reduce the induced voltage by as much as 40 per cent at a fre-
quency of 60 cycles. As bearing on the prevention of electric shock
from induced voltages on telephone lines, shielding has a disadvantage
in that it may, depending somewhat on the method of construction,
add to the chance of a lineman making contact with grounded metal.
Use of Cable. — A metallic sheath enclosing the conductors of a
cable is a type of shielding. The lead sheath of a 2% in. diameter aerial
telephone cable, if effectively grounded at the ends, as when directly
connected to an underground cable sheath, reduces the voltages in-
duced in the conductors within the cable by about 50 per cent at 60
cycles. The additional shielding brought about by the surrounding
earth when such a cable is placed underground is negligible at low
frequencies, although underground construction has an advantage in
affording a low-resistance ground for the sheath. The large number of
conductors in a cable afford mutual shielding which varies from a negli-
gible to a considerable amount depending upon many factors, impor-
tant among which is the extent of the cable beyond the ends of the ex-
posure. If two or more cables are close to one another through an
exposure, each benefits by the shielding action of the others, so that the
shielding increases with the number of cables.
If the lead sheath of the cable is surrounded by magnetic material
as by armoring or placing cable in iron pipe, the shielding may be
largely increased. With the form of iron tape armored cable referred
to in the Harrison-Silver paper, which is now in trial use, shielding at
60 cycles is about 80 per cent, assuming effective grounding. Armoring
a cable increases its cost substantially but has an advantage apart from
shielding in that the cable being protected by the armor against mech-
anical injury may be buried directly in the earth without conduit.
The armor is protected by impregnated wrappings but its life has yet
to be determined. The shielding afforded by this type of cable has been
studied experimentally under practical field conditions. Other instal-
lations and studies have been made abroad. It is probable that there
JOIXT DEVELOPMENT AND LOW-FREQUENCY INDUCTION 225
may be a field of use for this type of cable in situations for which it is
best adapted.
Coordinated Location of Lines. — Since the magnitude of induced
voltages for given power line conditions depends upon the inductive
coupling of the two classes of lines, which in turn is dependent upon
their relative location, particularly their separation and length of
parallelism, it is possible by advance cooperative planning of new
power and telephone line locations to minimize and in some cases to
forestall inductive effects in the telephone system. If the cost of
remedial measures which inductive exposures would render necessary
can be avoided, additional expense in locating lines to avoid such ex-
posures may be justified and where a complete solution is obtained in
this way both parties secure greater freedom in the construction and
operation of their lines. However, with the rapid expansion of both
services, the possibilities of complete solution by separation of lines
alone are becoming more and more rare, particularly for lines along
highways.
Coordination of Grounding Practises. — The occurrence of a fault on a
power system usually results in raising the ground potential at the
points of grounding as well as at the point of fault, but if steps are
taken to coordinate the grounding of the power system and the tele-
phone system serving the power company, particularly at transformer
and generating stations, the effects in the telephone system of the earth
potential gradient caused by a power fault may be minimized. For
example, if in a switching station the same ground should be used for
the power system neutral and for the telephone system, a power fault
might cause the switching station ground to rise many volts above the
distant telephone exchange ground, and result in operating the tele-
phone protectors and possibly interrupting service. If, however, in-
dependent grounds sufficiently separated are used at the switching
station, or an insulating transformer is placed in the telephone circuit,
the power neutral ground may rise in potential without unduly affect-
ing the telephone system.
Comprehensive consideration of the low-frequency coordination
problem involves a study of the reactions between the grounding
practises employed by power companies and those employed in tele-
phone and telegraph systems. There is considerable diversity in
practise with respect to methods of grounding. Some power trans-
mission lines and primary distribution lines are not provided with any
designed grounds, although most such lines have grounded neutrals and
a few lines are grounded in such a way that operating current flows
through the earth. In built-up communities there are underground
226 BFXL SYSTEM TECHNICAL JOURNAL
pipes, cables, and other structures along which current in the earth
will flow to a greater or less extent. These structures have varying
degrees of conductivity and some of them have, either by design or by
accident, high resistance joints. Consequently the paths of earth
currents are exceedingly complex. The conditions as to earth currents
and earth potentials necessary to be known in order to work out any
coordinated scheme of grounding would usually have to be determined
by tests.
The different kinds of grounds to be considered include those on:
power transmission circuit neutrals, lightning arresters, power distribu-
tion primary neutrals, power distribution secondaries, railway systems,
building conduits, telephone protectors, batteries, ringers, telegraph
circuits, lightning rods, electrolysis protection systems, various types
of signal circuits such as fire and police alarm systems, and so on.
The grounding practises for all these different systems should be care-
fully studied and coordinated in order to prevent so far as possible
harmful reactions among them. Such a study of course goes consid-
erably beyond the scope of this subcommittee.
Comparison of Different Protective Measures. — The ideal protective
measure would be one which furnished adequate protection and had
no unfavorable reaction from an economic or service standpoint on
the system to which it is applied. However, the work thus far has
not disclosed any measure which fully meets this ideal.
The relative advantage of different measures resolves itself into a
question of the best technical results which can be obtained at the
least over-all cost. The solution of problems consists of finding meas-
ures which afford the highest degree of protection which is practicable
and reasonable under the circumstances. In the investigation of a
specific case it may be found that certain protective measures can be
combined with other work in such manner that the cost is not wholly
chargeable to coordination for the reason that other results of value
are secured. For example, shielding may be obtained at small cost
if improvement of performance of a transmission line justifies the in-
stallation of ground wires; or, the benefits of shorter duration of in-
duced voltage by the use of high speed circuit breakers and high speed
relays may be secured in connection with a program for improving the
stability of power systems.
No other measure affords such complete protection against all effects
of induction as adequate separation. However, measures applied to
power systems such as fault current limitation which strike directly at
the source of low-frequency induction are of a basic character and per-
mit a closer association of the two classes of lines, a very important
JOINT DEVELOPMENT AND LOW-FREQUENCY INDUCTION 227
consideration in congested areas. Measures which affect only the
frequency of occurrence of faults, or their duration, while very helfpul,
are not as effective from a protection standpoint as measures which
limit the magnitude of residual currents and voltages.
As to measures which would allow telephone circuits to operate
through a strong inductive field, the use of lead-sheathed cable sur-
rounded by magnetic material seems to offer the physical possibility
of affording the most effective protection. Precautions would be re-
quired, however, to prevent the shielding structure itself from rising to
a dangerous potential with respect to earth. On open-wire lines where
the occurrence of high induced voltages cannot be prevented, some form
of protector for limiting the magnitude of voltage to ground seems to be
a logical line of development.
Devices such as acoustic shock reducers, which protect only against
a single effect of induced voltages, do not afford a solution of most
specific situations, but have to be used in combination with other
protective measures. In many situations, no single protective measure
is adequate and if the exposure is severe several may be required.
In considering the effects which a new exposure may produce, all
the relevant factors are capable of advance determination except
frequency of occurrence of induced voltages, which has to be estimated
on the basis of experience or judgment and a statistical analysis of line
failures.
Selection of measures to be employed in specific cases should be
made with the above considerations in mind to the end that the best
engineering solution may be obtained irrespective of whether the pro-
tective measures are applied to the telephone system, to the power
system, or to both.
Reaction of Physical Exposures and Lightning on Loio- Frequency
Induction Problem. — As telephone circuits which are exposed to in-
duced voltages may also be exposed to possible contact with power
circuits and to lightning, any comprehensive scheme of protection
must take into consideration the high currents resulting from contact
and the high voltage due to lightning. In this connection there are
some points of difference in the reactions on the protection scheme of
induction, contact, and lightning.
Contacts between power and telephone wires may occur at crossings
or conflicts or they may occur on joint pole construction as described
in the paper by Messrs. Huber and Martin. In any event such con-
tacts can occur only where the two lines are in close proximity, whereas
in cases of inductive exposure, a fault outside as well as inside the ex-
posure, may produce disturbances in the telephone circuits. Moreover,
228 BELL SYSTEM TECHNICAL JOURNAL
in cases of contact, wire or structure failures are generally involved while
faults may cause induction which do not involve falling wires. Con-
tacts impose on the telephone line the full voltage to ground of the
power conductor at that point, whereas induced voltage is usually only
a fraction of the power circuit voltage. This does not mean that the
imposed voltages due to contact are always higher than those due to
induction, because the majority of exposures to contact do not involve
the higher voltage circuits while the opposite is true regarding inductive
exposures. In cases of contact only part of the wires of the telephone
line are usually involved whereas in the majority of induction cases sub-
stantially the same voltage is induced on all the wires. The voltages
imposed on a telephone line by contact as well as those by induction
may extend over the full length of the conductors involved.
In addition to the effects of contact between wires of the two systems,
there is a distinct class of hazard to linemen of both utilities introduced
by situations of insufficient clearance due to improper construction or
inadequate maintenance on the part of one or both utilities.
Voltages on telephone lines by lightning produce effects somewhat
similar to the effects produced by power lines but lightning voltages
differ from the other voltages in that their duration is much shorter.
Lightning makes necessary protector discharge gaps of very high speed
of operation in order to prevent serious over-voltages on the telephone
system, whereas contacts with power circuits make necessary a pro-
tector of high current-carrying capacity.
Committee's Program of Work
The program of work on low-frequency induction undertaken by the
Joint Subcommittee on Development and Research through its project
committees is laid out to develop the essential facts bearing on the
problem of telephone protection in a broad sense, including causes,
effects, and remedial measures. The program covers not only the
technical but also the economic aspects of the problem. The problems
of lightning and physical contact under conditions of conflict or joint
use are also included, as the measures finally adopted must protect
against voltages from these sources as well as voltages induced by
power systems.
Extensive field trials of all promising protective measures, are under
way in order to determine their practicability under operating condi-
tions. As the work progresses, it is expected to issue from time to
time reports covering the applicability, efficacy, limitations, and condi-
tions of use, of various measures. This should result in a better under-
standing of the problem and more effective and economical solutions of
specific situations as they arise.
JOIXT DEVELOPMENT AND LOW FREQUENCY INDUCTION 229
Bibliography
Many features of power system design, operation and stability, as well as many
features of the telephone system, have a bearing on low-frequency mduction. It,
therefore has appeared impracticable to include a complete bibliography but the
following' references include the more important reports and articles on this subject.
1 "Inductive Interference between Electric Power and Communication Circuits.
Selected Reports of the Joint Committee on Inductive Interference, Published
by the Railroad Commission of the State of California, April 1, 1919.
The references in this volume of reports of greatest interest in connection
with low-frequency induction are:
"Balanced and Residual Voltages and Currents, pp. 34 and 119.
Technical Report No. 51— "Residual Voltage Due to the Line Unbalance
of Power Circuits Isolated from Ground— Effect of Circuit Configuration
Transpositions and Frequency," pp. 266-352. , ^ r.i
Technical Report No. 52— " Residuals Produced by a Ground on One Phase
of a Normally Isolated Three-Phase System." With supplemental
memorandum, pp. 353-376. ^ , • , r. n i
Technical Report No. 64 — "Computation of Induction between Parallel
Power and Communication Circuits," pp. 638-672.
Technical Report No. 65— " Coefficients of Induction for Communication
Circuits Paralleled by Three-Phase Power Circuits. Variation with
Relative Position and Configuration," pp. 673-1016.
Technical Report No. 68— "Effect of Protective Ground Wires of Power
Lines on Induction in Parallel Communication Circuits," pp. 1088-1093.
Technical Report No. 69— "Relation of Currents in Terminal Apparatus of
Telegraph Circuits to Induced Voltages and Location of Parallel," pp.
1094-1101. . ,^, . ,. , ^ . . ,
2 " Reports of Joint General Committee of National Electric Light Association and
Bell Telephone System on Physical Relations between Electrical Supply and
Signal Systems," edition of Dec. 9, 1922.
3 "Engineering Reports of the Joint Subcommittee on Development and Re-
search," National Electric Light Association and Bell Telephone System-
Report No. 4 — "An Investigation of Ground Faults on a 33-kv. Transmission
System and the Resulting Voltages in a Parallel Telephone System,"
May 29, 1929, pp. 7-47. , t , • t
Report No 5— "Athenia- Passaic Ground Potential and Induction Investiga-
tion," May 29, 1929, pp. 49-59. . , , •
4 S Kudo and S. Bekku— "The Transient Electromagnetic Induction on the
' Communication Line Caused by the Parallel Power Line," Researches of the
Electrotechnical Laboratory, No. 121, Tokyo— Nov. 1922. _ „^^ , ,
5. "Power Circuit Interference with Telegraphs and Telephones. S. C. Bartholo-
mew, with bibliography, ^. /.£.£. Jowrwa/, Oct. 1924. . , „,
6 "Power Distribution and Telephone Circuits— Inductive and Physical Rela-
tions," H. M. Trueblood and D. I. Cone, also discussion, A. I. E. E. Trans.,
Vol 44, 1925, pp. 1052-1064.
7 "Mutual Impedances of Grounded Circuits," G. A. Campbell, Bell System
Technical Journal, Oct. 1923, Vol. 2, pp. 1-30. , , ^ t^- r u, •
8 " Ober das Feld einer unendlich langen Wechselstromdurchflossenen Eintachleit-
ung " F. PoUaczek, Elektrische Nachrichten Technik, Sept. 3, 1926, pp. 339-359;
Jan.' 4, 1927, pp. 18-30. „ j j, n
9 "Wave Propagation in Overhead Wires with Ground Return, J. R. Carson,
Bell System Technical Journal, Oct. 1926, Vol. V, pp. 539-554. , ,, ,,
10 "Theory' of the Conduction of Alternating Currents through the Earth, G.
Haberland, Zeit. fur Angew. und Meek. 6, Oct. 1926, pp. 366-379.
11 "Ground Return Impedance— Underground Wire with Earth Return, J. R.
Carson, Bell System Technical Journal, \929,\o\.V\\\,f>p.9'i:-9&.
12 "Mutual Impedances of Ground Return Circuits," A. E. Bowen and C. L.
Gilkeson, A. I.E. E. Journal, Aug. 1930, p. 657. . „ , ,
13. " Method of Symmetrical Coordinates Applied to the Solution of Polyphase Net-
works," C. L. Fortescue, A. I. E. E. Trans., 1918, Vol. 37, pp. 1027-1115^
14 "Analytical Solution of Networks," R. D. Evans, Electric Journal, April, 1924
Vol. 21, pp. 149-154, and May 1924, VoL 21, pp. 207-213.
230 BELL SYSTEM TECHNICAL JOURNAL
15. "Equivalent Single-Phase Networks for Calculating Short-Circuit Currents Due
to Grounds on Three-Phase Star Grounded Systems," R. A. Shetzline,
A. I. E. E. Trans., 1924, \'ol. 43, pp. 875-883.
16. "Calculation of Short-Circuit Ground Currents on Three-Phase Power Networks,
Using the Method of Symmetrical Components," S. Bekku, G. E. Review,
Vol. 28, July 1925, pp. 472-478.
17. "Calculation of Sin^de-Phase Short Circuits by the Method of Symmetrical Com-
ponents," A. P. Mackerras, G. E. Rei'ie^v, Vol. 29, April 1926, pp. 218-231;
July 1926, pp. 468-481.
18. "Characteristics of Ground Faults on Three-Phase Systems," S. B. Griscom,
Electric Journal, Vol. 24, April 1927, pp. 151-156.
19. "Transmission Line Engineering," W. W. Lewis, McGraw-Hill Book Company,
1928, particularly chapters VI, VIII and X.
20. "Symmetrical Components." C. F. Wagner and R. D. Evans, Electric Journal,
Part I, Mar. 1928, p. 151; Part II, April 1928, p. 194; Part III, June 1928, p.
307; Part IV, July 1928, p. 359; Part V, Sept. 1929, p. 425; Part VI, Dec. 1929,
p. 571.
21. "Finding Single- Phase Short-Circuit Currents on Calculating Boards," R. D.
Evans, Elect. World, Vol. 85, April 11, 1925, pp. 760-765.
22. "An Alternating-Current Calculating Board," H. A. Travers and W. W. Parker,
Electric Journal, May 1930, Vol. 27, p. 266.
23. "The M. I. T. Network Analyzer — Design and Application to Power System
Problems," H. L. Hazen, O. R. Schurig and M. F. Gardner, A. I. E. E. Trans.,
July 1930, Vol. 49, p. 1102.
24. " Unterdriickung des Aussetzenden Erdschlusses durch Null Widerstande und
Funken Ableiter," \V. Petersen, E. T. Z., August 1918.
25. "Die Begranzung des Erdschlusstromes und die Unterdruckung des Erdschluss-
lichtbogens durch die Erdschlusspule," W. Petersen, E.T.Z., January 1919.
26. "The Petersen Earth Coil," R. N. Conwell and R. D. Evans, A. I. E. E. Trans.,
1922, Vol. 41, pp. 77-93.
27. "The Relation of the Petersen System of Grounding Power Networks to In-
ductive Effects in Neighboring Communication Circuits," H. M. Trueblood,
Bell System Technical Journal, 1922, Vol. I, pp 39-59.
28. "Arcing Grounds and Effect of Neutral Grounding Impedance," [. E. Clem,
A. LE. E. Trans., 1930, Vol. 49, No. 3, pp. 970-989.
29. "Grounding Banks of Transformers with Neutral Impedance and the Resultant
Transient Conditions in the Windings," F. J. Vogel and J. K. Hodnette,
/I. /.£.£. /owrwa/, October 1930, pp. 838-841.
30. "Uber die Schutzwirkung des Kabelmantels bel Induktionsbeelnflussungen von
Schwachstromkabeladern durch Starkstromleltungen," G. Grause and A.
Zastrow, WissenschaftlicJw Veroffentlichungen aus dem Siemens-Konzern,
Germany, 1922, Vol. 2, pp. 422H135, Abstract In Elektrotech. u. Maschinen-
bau, 2/4/23, Vol. 41, p. 95.
31. "The Pupin Cable Along the Electric Railwav Line — Schopfheim-Saeckingen,"
W. Rihl, Siemens-Schuckert Review, 1927, Vol. Ill, p. 169.
32. American Committee on Inductive Coordination — "Bibliography on Inductive
Coordination," Published by the American Committee on Inductive Coordina-
tion, Jan. 1, 1925.
status of Cooperative Work on Joint Use of Poles *
By J. C. MARTIN and H. L. HUBER
Because of the necessity of reaching the same customers, electric supply
and telephone lines commonly use the same streets and highways. In urban
communities, the joint use of poles for these two services has been very
widely adopted and practises for joint use construction have been established
from experience gained in past years. In rural communities, joint use is not
always practicable or economical. Joint use involves rnany engineering and
economic problems which have received the careful consideration of the Joint
General Committee of the National Electric Light Association and Bell
Telephone System.
This paper describes some of the problems which have been encountered
in joint use, and briefly outlines the work which is being conducted by the
Joint General Committee in connection therewith.
It is concluded that in specific cases proposed for joint use all factors
should be studied cooperatively by the companies concerned and that every-
thing practicable should be done to facilitate joint use construction and
extend its usefulness.
TELEPHONE and electric light and power services are supplied
in the same areas and to customers who are to a large extent
common to both utilities. It is therefore necessary that both types of
service be carried along the same streets and highways.
Experience has shown that safer and more satisfactory conditions
can often be secured if the power and telephone circuits are carried
on the same poles. This is due in part to the fact that clearances and
climbing space can be more readily maintained where both classes of
circuit are carried on the same poles rather than on separate poles on
the same side of the street. Where separate lines are placed on oppo-
site sides of the streets and alleys, it is difficult to secure and maintain
proper clearances for service wires to buildings where these cross the
line of the other utility.
Joint use of poles by the power and telephone companies results in
the use of fewer poles on streets and highways and better appearance
of aerial lines. It is, therefore, more desirable from the public point of
view. It conserves pole timber and in many cases is more economical
to both classes of utility than separate lines.
Because of the above mentioned advantages, joint use of poles by
power and telephone companies has been widely adopted. No com-
plete data are available as to the extent of such joint use at the present
time, but it is estimated that there are at least five million poles jointly
used by the power and telephone companies in the United States.
* Part IV^ of the Symposium on Coordination of Power and Telephone Plant.
Presented at the Winter Convention of the A. I.E. E., New York, N. Y., January
26-3U. 1931. Published in abridged form in Ekclrical Engineering, March, 1931.
231
232 BELL SYSTEM TECHNICAL JOURNAL
Both of these classes of utility have been growing rapidly in the past
twenty-five years and the development, design, and construction of the
physical plant of each has kept pace with the growth in territory and
number of customers served.
While earlier types of distribution plant were such that the possibility
of contacts between wires of the two utilities and other hazards could be
satisfactorily met by proper construction methods, protective devices,
etc., later developments have increased the use of types of power distri-
bution circuits regarding which questions frequently arise as to how
service can be properly maintained and extended on jointly used poles.
These questions have received and are receiving careful considera-
tion by the Joint General Committee of the National Electric Light
Association and Bell Telephone System. This committee has recom-
mended certain principles and practises for the joint use of wood poles
which are intended for use as a basis on which electric supply companies
and communication companies should work out their mutual problems
and has undertaken important research work in connection with these
matters through its Joint Subcommittee on Development and Re-
search.
The principles and practises mentioned were presented in a report
of the Joint General Committee under date of February 15, 1926, and
while it is beyond the scope of this paper to consider these principles in
detail, the following recommendations are of interest in that they
indicate the way in which this matter is generally being approached:
Each party should:
(a) Be the judge of the quality and requirements of its own service,
including the character and design of its own facilities, both now
and in the future.
(b) Determine the character of its own circuits and structures to be
placed or continued in joint use, and determine the character of
the circuits and structures of others with which it will enter into
or continue in joint use.
(c) Cooperate with the other party so that in carrying out the foregoing
duties, proper consideration will be given to the mutual prob-
lems which may arise and so that the parties can jointly deter-
mine the best engineering solution in situations where the
facilities of both are involved.
It will be observed that while each party retains full responsibility
for facing and meeting its own problems, it is recommended that both
parties cooperate in working out mutual problems involving the joint
use of poles and in finding the best over-all engineering solution in
each situation. These are among the most important of the principles
COOPERATIVE WORK ON JOINT USE OF POLES 233
which have been recommended iind are the basis upon which practically
all cooperative work is being carried forward.
It is the purpose of the following paragraphs to describe what has
been done and what is being done by the Joint Subcommittee on De-
velopment and Research in connection with the engineering and econ-
omic problems which have been encountered in joint use work.
Construction Practises
Joint use construction practises have undergone almost continual
change and improvement from the time joint use was first adopted and
continued development is to be expected in the future. However,
many of the fundamental requirements for securing satisfactory con-
ditions on jointly used poles were recognized at an early date and form
the basis for present day practise.
In the matter of relative levels it has been recognized that power
wires should as far as practicable be carried in the upper position. In
general, they are larger and stronger than the telephone wires. This
is inherently so because of the current carrying capacity required.
Placing power wires in the upper position on jointly used poles avoids
the necessity of telephone linemen climbing through power circuits, the
exact nature and characteristics of which they are not always familiar
with.
Clearances must be provided which give sufficient space below the
power wires so that power linemen will not have to come in contact
with telephone wires while they are working on power wires. This
neutral space must also provide sufficient clearances above the tele-
phone equipment so that telephone linemen may work on the telephone
plant without danger of coming in contact with power equipment.
Clear climbing space must also be provided so that linemen may climb
poles without having to be extremely careful to avoid falls or contacts
with circuits from which they may receive physical injuries.
Fig. 1 shows one method for securing satisfactory conditions on a
jointly used pole carrying circuits which both the power and telephone
groups have recognized as being suitable for joint use.
In the matter of mechanical strength, joint use follows the practise
in the construction of separate lines. That is, strength of construction
should be provided such as to stand, with reasonable factors of safety,
storm conditions which experience indicates are likely to occur from
time to time in any particular area.
With regard to the matter of insulation and electrical strength, prac-
tises as to the size and type of power insulators hiive followed develop-
ments in the general field of power construction. Wires to street lights
234
BELL SYSTEM TECHNICAL JOURNAL
and underground connections to aerial plant require vertically run
wires on jointly used poles. The location, insulation and mechanical
protection of these have received special consideration to eliminate
hazards to workmen.
Sufficient clearances between vertically run circuits of one type and
the equipment of another utility on jointly used poles have also been
a a
Communication
Cable Terminal
Fig. 1 — Typical jointly used pole.
found to be very important from the standpoint of avoiding interrup-
tion to power and telephone services.
In the course of electrical storms, lightning may induce high voltages
on either supply or communication wires. If the separation between
the supply and communication facilities is not adequate at any point,
these induced voltages may break down the insulation and arc between
the two as illustrated in Fig. 2. Damaged plant may, of course, result
from lightning alone. However, when lightning has established an arc
between the power and communication circuits the normal voltage of
COOPERATIVE WORK OX JOINT USE OF POLES 235
the supply circuit may maintain the arc. This results in the transfer
of power into the telephone plant at voltages which may be well above
that for which it is insulated and may cause trouble on both the power
and telephone system. This sort of abnormal belongs to the general
class that includes insulator flashovers, short circuits on cables, tree
grounds and similar power system occurrences that always carry the
probability of damage to the power system or service.
While vertically run attachments with improper clearances have
played a large part in causing such occurrences, any situation where
fc-*
Fig. 2 — Frayed insulation showint^ breakdown of insulation between power and
telephone plant.*
insufftcient clearance between power and telephone facilities is pro-
vided may result in similar trouble.
Emphasis has, therefore, been placed in present day standards on the
necessity of maintaining proper clearances as well as strength of con-
struction to prevent this kind of abnormal. Experience has shown that
where these clearances are adhered to this type of abnormal is kept to a
reasonable minimum.
The Joint General Committee is giving careful consideration to the
matter of construction standards on jointly used poles. Pending the
development of complete specifications covering recommended prac-
* In order to obtain a satisfactory photograph of the points of arc the vertical
drop has been straightened out so that the clearance shown is much greater than
existed at the time of the arc.
236 BRLL SYSTEM TECHNICAL JOURNAL
tises under various conditions, they have recommended the National
Electrical Safety Code to be used as a guide to practise.
Protective Devices
Both telephone and supply circuits are eciuipped with protective
de\ices which arc fundamentally the same in principle. They may he
divided into two general classes:
1. Those which provide protection from abnormal voltages consisting
of protector blocks in the telephone plant and lightning arresters
in the supply system.
2. Those which provide protection from abnormal currents consisting
of heat coils and fuses in the telephone plant and fuses and cir-
cuit breakers in the supply systems.
These protective devices are a secondary defense against abnormal
conditions which it is impracticable to avoid either by design or through
adherence to construction standards.
P2ven when all practicable precautions with regard to clearances,
strength of construction and insulation have been taken, accidental
breaks occur in both power and telephone wires. In some cases there
are direct contacts between such wires. Higher than normal potentials
are also introduced into the telephone and power circuits by lightning
and other causes.
It is because of the limitations of protective devices and other pro-
tective measures that joint use with certain types of circuits has been
in question. Considerable differences of opinion exist between engi-
neers as to the degree of hazard involved in joint use between telephone
plant and power circuits of various types, voltages, and connected
power. The problem has increased in importance as the use of higher
distribution voltages and greater generating capacity have been em-
ployed.
This matter is under investigation by the Joint Subcommittee on
Development and Research. Studies are now in progress in one rural
area and in one suburban area to determine the over-all advantages and
disadvantages of the use of higher distribution voltages and of joint
use with these voltages under present conditions.
The first experimental work done by the Joint Subcommittee in
connection with these problems was a detailed study of the character-
istics of various types of fuses. This study covered all of the well
known commercial types of telephone fuses and a number of experi-
mental models. The operating characteristics of these fuses were
obtained at voltages of 2300, 4000, 7500 and 13,200. The current
range was from 16 to 1000 amperes. These tests were carried on in a
COOPERATIVE WORK ON JOINT USE OF POLES 237
laboratory where 20,000 kv-a. of generating capacity was available.
The tests showed the dependability that could be placed upon the
various fuses for interrupting voltages of the range from 2300 to 13,200
volts. They show^ed under what conditions the fuses could be de-
pended upon and the ranges where the available type of fuses could not
be depended upon for safe operation.
A number of the experimental models showed considerable promise
of improvement over existing models, and this work will be carried
further to determine what improvements can be made in the operating
characteristics of fuses.
The next phase of the problem taken up included a study of the
operating characteristics of various types of overvoltage protectors
suitable for use on communication circuits. The experimental work
covered breakdown with direct current, 60-cycle alternating current
and a complete study with a cathode ray oscillograph of the behavior
under steep wave fronts for carbon block protectors, neon and vacuum
tubes.
These tests showed that the carbon block protector has a breakdown
point with all types of applied wave fronts which is sufficiently fast
and low to protect the insulation that is now used in the communica-
tion plant, as shown by similar tests on condenser and cable paper.
The shortcomings of these blocks lie in their tendency to permanently
ground the circuit when carrying current for any appreciable length of
time.
The tests with steep wave fronts were carried to a rate of rise of
36,500 volts per microsecond, and it was determined by tests of
propagation of steep wave front voltages through telephone cable that
it was practically impossible to subject the plant to voltages with any
faster rate of rise than those used in the protector tests.
The problem of adequate protection of the telephone plant in joint
use, obviously, cannot be solved by the development of the telephone
protective devices alone. The protective devices in the power system
are equally important.
One of the important functions of the power system protective de-
vices is that of clearing power system faults in a reasonable time inter-
val. Obviously, telephone protective equipment cannot be expected
to prevent damage to telephone plant in case of contact between the
wire circuits of the two utilities when power system protective devices
fail to operate and the physical contact of the circuits is maintained over
an indefinite period of time.
One problem in the development and research work is the fixing of
the part that the protective devices on each system must play in abnor-
238 BELL SYSTEM TECHNICAL JOURNAL
mal conditions. It is necessary that the over-all protective equipment
be adequate and that the burden of overcoming weaknesses in the
protective equipment of one system be not thrown on the protective
equipment of the other. There are inherent limitations in both classes
of protective equipment that must be defined.
Therefore, the next step in this investigation is a determination of
the over-all characteristics of power circuit and telephone circuit pro-
tection under typical conditions of contact between the two plants.
While protective devices are an important element in connection
with joint use involving certain types of power circuits employing the
higher distribution voltages, there are also other important considera-
tions. The general insulation of the telephone plant must also be
considered, especially in connection with drop loops attached to and
entering subscribers' premises. These matters are also being studied
by the Joint Subcommittee.
All of these problems, as is the case of others being studied by the
Joint General Committee, are being approached on the basis of deter-
mining the best over-all engineering solution such that both systems can
provide their services in the most convenient and economical manner.
Inductive Coordination
In the early history of joint use, noise induction problems involving
street lighting circuits appeared. Other interesting problems were
encountered such, for example, as the accidental grounding of one
corner of an isolated delta power system with its resulting unbalanced
voltage inductive effects on open-wire telephone circuits, which type
of telephone construction then predominated.
As these problems arose they received careful study and with the
development and extended use of telephone cables and the use of
improved operating methods in power and telephone distribution
generally, inductive coordination of power and telephone distribution
systems in the urban communities became less troublesome and did
not for a time receive any large amount of consideration.
However, during recent years the introduction and extended use of
various types of multi-grounded distribution systems described in the
paper by Messrs. Harrison and Silver and the existence of certain types
of signaling on local telephone circuits, have contributed toward mak-
ing important the consideration of noise inductive effects in connection
with joint use. This matter is discussed more fully in the paper by
Messrs. Harrison and Silver.
The technical factors involved in inductive coordination problems
under joint use conditions are complicated. The details regarding
COOPERATIVE WORK ON JOINT USE OF POLES 239
these factors and the results of the extensive studies of these matters
by the Joint Subcommittee on Development and Research are described
in the paper by Messrs. Wills and Blackwell.
The various operating problems which have arisen almost since the
birth of the power and telephone industries and the investigations con-
ducted by the Joint Subcommittee on Development and Research
indicate the importance of giving careful consideration to the inductive
coordination features of joint use and of including this factor in studies
of the relative advantages and disadvantages of joint use as compared
with separate lines. This factor should, of course, be considered from
both its technical and economic aspects.
Much can be accomplished in the inductive coordination of the two
distributing systems by cooperative advance planning. In urban
areas where the telephone circuits are largely in cable, there is about
a two to one ratio in the inductive effects between a joint line and sepa-
rate lines across the street. In rural areas where the telephone circuits
are largely open wire, the ratio of the inductive effects on joint lines
as compared with separate lines across the highway, is much greater,
other things being equal.
In urban areas the power and telephone companies can through
cooperative planning frequently arrange to establish important power
feeders and telephone circuits on separate streets and thereby avoid
large inductive effects and permit more extensive joint use of branch
lines. A careful review of the equipment used on the powder and tele-
phone circuits and the introduction of operating practises designed to
limit the inductive susceptiveness of the telephone circuits and the
inductive influence of power circuits, form an important part of ad-
vance planning and cooperation.
As described in the paper by Messrs. Wills and Blackwell, these
latter factors include such items as limitation of the odd triple fre-
quency series arising in Y-connected generators feeding directly on the
line and in single-phase service transformers. Suitable limitations
of the unbalances existing among the loads connected between the
three-phase conductors and the neutral, limit the ground return com-
ponents.
Grounding of aerial telephone cable sheaths to provide for increased
shielding and the use of central office and station equipment providing
a higher degree of balance with respect to ground are helpful.
The matter of joint use may involve both rural and urban commun-
ities. It is more generally associated with the latter because of the
severe limitations in physical space available for utility use. In the
case of rural lines where the telephone circuits are largely in open wire
240 BELL SYSTEM TECHNICAL JOURNAL
and the exposures between particular circuits are likely to be long, joint
use is not always practicable. In these cases locations for separate
lines are usually available.
Furthermore, joint use in rural areas is not always economical from
a purely construction standpoint due to the fact that relatively longer
spans can often be used on the power lines and both utilites are able
in many instances to use shorter and lighter poles than would be
practicable in joint use. Joint use with telephone toll circuits or
power transmission lines has not, in general, been found desirable.
Types of construction vary so widely and service requirements and
inductive effects are such that it becomes uneconomical to carry out
such construction.
Conclusions
Joint use of poles by power and telephone companies has many
advantages, both from the standpoint of the public and from the
standpoint of the wire using companies. This is especially true in
built-up communities.
Important problems brought about by developments in practises,
particularly in the use of high voltage distribution, remain to be solved.
Careful adherence to generally accepted practises with regard to
clearances, strength of construction, insulation and inductive coordin-
ation is necessary in order that the advantages of joint use can be
secured.
In considering specific cases proposed for joint use, it is advisable that
all of these factors be studied cooperatively by the companies con-
cerned, to the end that good service, safety and economy by both classes
of utility may be promoted.
It is important that everything practicable should be done to facili-
tate joint use construction and extend its usefulness. The Joint
General Committee of the National Electric Light Association and
Bell Telephone System is continuing its efforts in this connection.
Symposium on Coordination of Power and Telephone Plant
Closing Remarks *
By B. GHERARDI
THE papers which have been presented here today bring out clearly
the progress which has been made by the power and telephone
companies in the study and development of methods for coordinating
their facilities. It seems to me that this is an outstanding example
of what can be accomplished through joint study and cooperative
methods generally.
This work illustrates also the way in which the field of activity of the
engineer is broadening. While the main duty of the engineer may still
be the application of physical laws to accomplish the most satisfactory
results in the most economical manner, the very organization of society
which has resulted from these applications of physical laws, requires
the engineer, if he is to play his full part, more and more to include in
his considerations the broad economic and human factors which govern
the success of social and business enterprises. In the work described
in this symposium the approach has been not only the consideration
of the complicated technical questions involved, but the working out
as well of these questions on the basis of good business relations be-
tween two large utilities, having in mind that both have the responsi-
bility for providing important services to the same public.
I would like to reiterate certain fundamentals which have played an
important part in bringing about the present satisfactory situation.
First, is that of getting together and getting acquainted, to the end
that frank and friendly discussions will be promoted and misunder-
standings avoided. Second, is the continued development of technical
information for the coordination of power and communication systems
adequate to keep pace with the rapid amplification and growth of
these two services. Third, is a desire on the part of the companies
concerned to work out each case in accordance with the best over-all
engineering solution as though both utilities were under the same man-
agement. Where there is such a desire, the working out of the job is
largely a matter of detail and results are assured which will be fair to
all concerned.
We feel that the results of the cooperative work have been good from
every point of view, and I want to express the appreciation of the Bell
* Presented at the Winter Convention of tiie A. I. E. E., New York, N. Y.,
January 26-30, 1931.
241
242 BELL SYSTEM TECHNICAL JOURNAL
System people of the broad spirit of coopereition in which this matter
has been approached by the power companies. I heartily join with
Mr. Pack in his feeling of satisfaction for having taken part in the initial
work which has brought about the present fine relations between the
power and telephone companies and the effective handling of the various
types of situation involving coordination.
Overseas Radio Extensions to Wire Telephone
Networks *
By LLOYD ESPENSCHIED and WILLIAM WILSON
The development of intercontinental telephony through the agency of
radio links connecting between the land networks is traced and its present
trends indicated. A description is given of the facilities employed by the
Bell System for overseas connections and connections to ships at sea. The
transmission results secured with these facilities are set forth and some pecul-
iar short-wave phenomena discussed. International problems of frequency
use and conservation are briefly summarized. A fairly comprehensive bib-
liography of technical papers on transoceanic telephony is included at the end
of the paper.
Introduction
THE progress which long-distance electric communication is
making in tying the world together is perhaps nowhere more
interestingly illustrated than in the developments which are now
taking place in the interconnection of widely separated wire telephone
networks by means of overseas radiotelephone links. It was only a
few years ago, in 1927, that telephone service was first extended across
the barrier of the North Atlantic and a beginning made in the inter-
connection of the great telephone networks of North America and of
Europe. Rapid progress has been made since then in the further de-
velopment of the North Atlantic facilities and in the extension of
radiotelephone links from these wire telephone networks outward in
other directions, until today such links span a large portion of the
globe.
Since it is the nature of telephony that the circuits are employed
personally by the telephone users it is necessary that these inter-
connecting links be of a high standard of transmission effectiveness
and be free from interference. Also it is important that they be
reliable in operation and continuously available during the operating
periods, for the usefulness of telephone service is in part dependent
upon its being immediately available on call. Although these re-
quirements are not yet being fully met, the circuits already in opera-
tion are very effective and are proving to be valuable additions to the
world's communication facilities.
The progress which is being made and the problems which are
arising in the establishment of these systems and in the coordination
* Presented before Fifth Annual Convention of the Institute of Radio Engineers,
August, 1930; Proceedings of 1. R. E., February, 1931.
243
244 BKLL SYSTEM TECHNICAL JOURNAL
of tliein into a world-wide telephone network appeal to the imagi-
nation and challenge the best efforts of communication engineers.
Especially is this development of interest to radio engineers since in
this pioneering stage the interconnecting links are being forged by
radio. W'ork is also going forward in the development of new types
of submarine telephone cables for this purpose and undoubtedly such
cables will in time play a large part in fortifying the more important
of the world routes. The radio part of the picture is, however, quite
enough in itself and this paper is, therefore, largely confined to this
phase of the subject.
There is given first, a sketch of the wire telephone networks and the
interconnecting links as they exist today, second, a picture of the
transmission results which are being obtained in the operation of
some of these overseas links, and finally, a discussion of the more
important phenomena and problems involved in the radio trans-
mitting medium.
The Existing World Telephone Picture
A simplified picture of the present telephone development of the
world is given in the map of Fig. 1. Only the principal areas of tele-
phone development are indicated, by the shaded portion, and only the
more important routes of the wire networks have been sketched in.
The figures give the approximate number of telephone subscribers in
each continental area.
It is, of course, these networks which give direct access to millions
of people in offtces and homes and permit of the personal contact
which characterizes telephone communication. It is natural, there-
fore, that they should be the foundation of the world-wide system
which is growing up. The larger of these networks already spread
over national boundaries so that the engineering problem is primarily
one of interconnecting the networks, generally comprising groups of
countries, rather than that of directly interconnecting by radio all of
the component countries. The points within each network at which
the interconnections are made may be expected to be determined
largely by considerations of traffic and of operating efficiency. The
differences of time and of languages between these w^idely separated
areas, and, of course, the expense of providing reliable interconnec-
tions over these distances, are factors which will naturally limit the
volume of use to be made of these connections. That they are des-
tined to fulfill a very real need is already proven, however, by the
services which are now being given.
/
>y
L
OVERSEAS RADIO EXTENSION 245
Development of Ixtercoxxecting Links
The present status of the development of these transoceanic radio-
telephone links is illustrated in Fig. 2. There are shown the circuits
which are in operation and also the projects which have been reported
as under consideration or under construction. These telephone paths
will be observed to correspond in general with the routes followed by
the ocean telegraph and radiotelegraph services, in fact with the trade
routes of the world, along which community of interest has been built
up. Thus a certain orderly arrangement of the services is being
realized naturally.
In general, there may be said to be five major groupings:
1. The North American-European connections. These are, of course,
of outstanding importance because of the economic and social
interest which exists between the two continents and because
they connect with the large telephone wire networks on both
sides of the Atlantic. North America and Europe combined
account for about 32 million telephones out of a world total of
about 35 million. The present situation on the North Atlantic
route is discussed later on.
2. North America-South America.
3. South America-Europe.
4. Europe to Africa, Asia, and Oceania. The connections to Africa
and to Oceania represent the interest which some of the Euro-
pean nations have in associated commonwealths and in colonies.
5. North America to Pacific points and the Far East. These are in
the construction and project stage.
Most of these services are being given on a part time basis although
that across the North Atlantic has been found to require 24-hour
service and that between North and South America is for the full
business day. Some of the circuits from Europe to South America
and to the East Indies are not yet connected fully into the wire tele-
phone network. The circuits which are in operation between South
America and Europe instead of connecting into the European network
by means of a single station are shared on a part time basis by several
stations located in different countries in Europe, as is indicated by
the forked lines in the figure.
One advantage of the use of radio for these services, particularly
in this pioneering stage during which traffic over many of the routes
is likely to be small, is the ability to share the use of a transmitting
channel as between a number of receiving points where wire lines are
not available. A representative case of this kind would be that of an
OVERSEAS RADIO EXTENSION 245
Development of Interconnecting Links
The present status of the development of these transoceanic radio-
telephone links is illustrated in Fig. 2. There are shown the circuits
which are in operation and also the projects which have been reported
as under consideration or under construction. These telephone paths
will be observed to correspond in general with the routes followed by
the ocean telegraph and radiotelegraph serv^ices, in fact with the trade
routes of the world, along which community of interest has been built
up. Thus a certain orderly arrangement of the services is being
realized naturally.
In general, there may be said to be five major groupings:
1. The North American-European connections. These are, of course,
of outstanding importance because of the economic and social
interest which exists between the two continents and because
they connect with the large telephone wire networks on both
sides of the Atlantic. North America and Europe combined
account for about 32 million telephones out of a world total of
about 35 million. The present situation on the North Atlantic
route is discussed later on.
2. North America-South America.
3. South America-Europe.
4. Europe to Africa, Asia, and Oceania. The connections to Africa
and to Oceania represent the interest which some of the Euro-
pean nations have in associated commonwealths and in colonies.
5. North America to Pacific points and the Far East. These are in
the construction and project stage.
Most of these services are being given on a part time basis although
that across the North Atlantic has been found to require 24-hour
service and that between North and South America is for the full
business day. Some of the circuits from Europe to South America
and to the East Indies are not yet connected fully into the wire tele-
phone network. The circuits which are in operation between South
America and Europe instead of connecting into the European network
by means of a single station are shared on a part time basis by several
stations located in different countries in Europe, as is indicated by
the forked lines in the figure.
One advantage of the use of radio for these services, particularly
in this pioneering stage during which traffic over many ot the routes
is likely to be small, is the ability to share the use of a transmitting
channel as between a number of receiving points where wire lines are
not available. A representative case of this kind would be that of an
246 BELL SYSTEM TECHNICAL JOURNAL
important central station linked with a continental wire network from
which it is desired to establish connections with a number of smaller
outlying points. This possibility is not as simple as it may appear,
however, because there enter the problems of directive antennas, of
shifting frequencies if widely different distances are involved, and of
not permitting the return transmission to be materially weaker than
the outgoing transmission which means the use of relatively powerful
stations at the outlying points. In general, these short-wave stations
represent rather large investments and in working out interconnecting
arrangements of this kind it is important to fit together the schedules
at the various stations so as to minimize lost circuit time and to avoid
leaving stations in idleness.
North Atlantic Facilities
Of the four circuits which now exist across the North Atlantic, as
indicated in Fig. 2, one is the long-wave circuit, with which the service
was originally started, and three are short-wave circuits. The dashed
line, shown in the figure, between New York and London indicates an
additional long-wave circuit which is planned. There is also indicated
in the figure the ship-to-shore telephone service on the North Atlantic
which connects with the land line network on either side.
The transatlantic long-wave system has already been the subject
of technical papers ^ and need not be described in detail. It operates
on a single side-band carrier suppression system in a frequency band
centering at 60 kc. The single side-band system is used to minimize
the frequency space occupied. The single band is used alternately for
transmission in the two directions by means of voice actuated switch-
ing devices at the New York and London terminals. For the purpose
of minimizing the principal limitation of long waves, that of "static,"
the receiving stations are located as far north as is reasonably possible
and use is made of directive receiving antennas.
The three short-wave circuits which have been provided on the
North Atlantic route add materially to the trafiic capacity but are
erratic in their behavior and their usefulness is dependent, in a large
measure, upon being operated in combination with the more stable
long-wave circuit. All three short-wave circuits are affected similarly
by the adverse conditions accompanying magnetic storms, whereas
long- wave transmission is not materially affected by these conditions
except at night.^ The second long-wave circuit is planned to provide
a more balanced combination of facilities as well as to add to the total
1 See attached bibliography.
- Bibliography 6, 14, 15.
WELLINGTON
TRArFC OPtllATORS POSITIOH
TECHNICAL OPERATORS POSITION
REaiVING . _
RELAY ^ 1
RECEIVING FILTERS
REPEATER FILTER BALANCED OEMOOUUTOR NO. 2
SINGLE SIDE BAND CARRIER RESUPPLIED RECEIVER
INT. FREQ.
SINGLE SIDE BAND CARRIER ELIMINATED TRANSMinER
BALANCED MODULATOR N0.1 fH-TEB BALANCED MODULATOR N0.2 /'l-TES ^ AMPLIFIER AMPLIFIER^
Fig. 4 — Schematic of printer
5 transatlantic radio telephoi
OVERSEAS RADIO EXTENSION 247
circuit capacity across the Atlantic. In this connection, it should be
noted also that a new type of submarine telephone cable is under
development and is planned to be laid across the North Atlantic when
completed. While this cable will provide only one two-way circuit,
it is expected to be free from atmospheric disturbances and to fortify
greatly the telephone service between North America and Europe.
The ship-shore telephone service which is being given on the North
Atlantic includes a land station connection with the land line network
in both the United States and in England and through these land
stations service is given to most of North America and Western
Europe. Four of the larger transatlantic vessels are equipped. The
service may be expected to include in time additional shore stations
and many other vessels. It is an example of a class of service for
which radio alone is available, that of extending telephone service to
moving craft at sea or in the air.
Short-Wave Technique
With the exception of the long-wave circuit across the North At-
lantic, all of the links indicated in Fig. 2 are of the short-wave type.
As to these different short-wave stations throughout the world, there
is, of course, considerable difference between them in the requirements
which are being met and the performance obtained. However, the
same fundamental principles are being followed in all of the countries
and the short-wave telephone technique may be said to be rather
remarkably alike throughout the world. Transmission is on the or-
dinary double side-band basis since the necessity for narrowing the
band is not of great importance in the present state of the art and the
difficulty of single side-band operated at high frequencies is very much
greater. In general, the transmitters are of the vacuum tube type
employing master oscillators which are stepped up in frequency and in
power for the final transmission ; directive antennas are employed for
both transmitting and receiving, and in the receiving apparatus use
is made of the double detection principle with its advantages in giving
stable operation with high amplification and high selectivity.
In the case of the radiotelephone stations which connect with the
United States, the short-wave technique is further characterized by
the use of transmitting sets which are provided with a piezo-crystal
oscillator with temperature control for stabilizing the transmitting
frequency, and the use of interchangeable coils which permit the fre-
quency of the transmitter to be changed in keeping with the require-
ments for the different times of the day and year. The carrier output
of 15 kw. corresponds to a peak output of about 60 kw. The hnal
OVERSEAS RADIO EXTENSION 247
circuit capacity across the Atlantic. In this connection, it should be
noted also that a new type of submarine telephone cable is under
development and is planned to be laid across the North Atlantic when
completed. While this cable will provide only one two-way circuit,
it is expected to be free from atmospheric disturbances and to fortify
greatly the telephone service between North America and Europe.
The ship-shore telephone service which is being given on the North
Atlantic includes a land station connection with the land line network
in both the United States and in England and through these land
stations service is given to most of North America and Western
Europe. Four of the larger transatlantic vessels are equipped. The
service may be expected to include in time additional shore stations
and many other vessels. It is an example of a class of service for
which radio alone is available, that of extending telephone service to
moving craft at sea or in the air.
Short-Wave Technique
With the exception of the long-wave circuit across the North At-
lantic, all of the links indicated in Fig. 2 are of the short-wave type.
As to these different short-wave stations throughout the world, there
is, of course, considerable difference between them in the requirements
which are being met and the performance obtained. However, the
same fundamental principles are being followed in all of the countries
and the short-wave telephone technique may be said to be rather
remarkably alike throughout the world. Transmission is on the or-
dinary double side-band basis since the necessity for narrowing the
band is not of great importance in the present state of the art and the
difficulty of single side-band operated at high frequencies is very much
greater. In general, the transmitters are of the vacuum tube type
employing master oscillators which are stepped up in frequency and in
power for the final transmission; directive antennas are employed for
both transmitting and receiving, and in the receiving apparatus use
is made of the double detection principle with its advantages in giving
stable operation with high amplification and high selectivity.
In the case of the radiotelephone stations which connect with the
United States, the short-wave technique is further characterized by
the use of transmitting sets which are provided with a piezo-crystal
oscillator with temperature control for stabilizing the transmitting
frequency, and the use of interchangeable coils which permit the fre-
quency of the transmitter to be changed in keeping with the require-
ments for the different times of the day and year. The carrier output
of 15 kw. corresponds to a peak output of about 60 kw. The final
248
BELL SYSTEM TECHNICAL JOURNAL
power stage of such a set is shown in Fig. 3. The units marked 1,2,
and 3 are the water jackets for three of the six double-ended, 10-kw.
tubes, the other three being on the other side of the mounting. The
circuit is of the push-pull type.
Fig. 3 — Short-wave radiotelephone transmitting center of the American Tele-
phone and Telegraph Company, Lawrenceville, X. J. Six lO-kw. tubes used in
one of the output stages of a transmitting set. Coupling coils on right, monitoring
amplifier boxes at lower right.
The radio receivers employed in the United States are built so as
to have low intrinsic noise and sufficient gain to enable very small
field strengths, of the order of 1 nv. per m., to be detected and raised
to the required telephone speech level. They are equipped with auto-
OVERSEAS RADIO EXTENSION 249
matic gain control which minimizes the fading variations in speech
volume. One of the radio receivers employed at the Netcong, N. J.,
receiving station is illustrated in Fig. 4. The antenna leads are
brought in beneath the floor in the concentric pipes which are seen to
Fig. 4— Short-wave radiotelephone receiving center of the American Telephone
and Telegraph Company, Netcong, X. J. Radio receiver for South American cir-
cuit. Antenna concentric pipe transmission lines enter set overhead.
rise at the right and connect with the input of the set on the upper
left-hand panel. The first two vertical bays are the radio set proper,
including the automatic gain control. The third bay, on the right,
includes the volume indicator and control and the line connecting
equipment.
250
BELL SYSTEM TECHNICAL JOURNAL
In general, three wavelengths are used, one around 19,000 kc. (16
meters), one around 14,000 kc. (21 meters) and one around 9,000 kc.
(3v? meters), and each transmitter and receiver is arranged so that it
can be connected at any time with a directive antenna designed for
each of these frequency ranges. The transmitter antenna gains are
about 17 db over a one-half wave antenna. These short-wave radio-
telephone facilities which connect the American telephone network
with Europe and South America have already been the subject of
technical papers ' and need not be described in further detail. An air
view of the Lawrenceville, N. J., transmitting station is given in
Fig. 5. The longer of the two lines of towers supports the antennas
Fig. 5 — ^Lawrenceville transmitting station. Aerial view — -South American an-
tenna in the foreground; European antenna in the background. Two buildings
each containing two transmitters are shown.
for the three short-wave circuits to England, and the shorter line of
towers the antennas for the single circuit to the Argentine. Some idea
of the magnitude of the plants emplo>ed for these short-wave circuits
may be had from this photograph. The longer line of antennas is
approximately one mile long, consisting of twenty-one 185-ft. towers.
Substantial fireproof buildings are provided for the transmitting sets
and auxiliary equipment. Probably every operating agency which has
^ See bibhography.
OVERSEAS RADIO EXTENSION 251
had experience with short-wave operation reaHzes that the cost of
such radio facihties is proportional to the standard of service and to
the degree of reHability and exactitude of operation which is under-
taken in the terminal stations.
JoiNiNCi OF A Radiotelephone Link with Wire Network
The manner of joining the transoceanic radio links with the wire
network to meet the requirements of through two-way transmission is
an interesting and important development in itself. In general this
technique is an outgrowth of wire telephone practice and is so new as
not yet to have been fully applied to all of the radiotelephone links in
existence.
The problem is that of how to form the two oppositely directed
speech channels which comprise the radiotelephone link itself into the
usual two-w^ay telephone circuit suitable for use as a regular telephone
toll line and for termination before long-distance traffic operators at
each end.
The transmission equivalent of the radio paths may be continu-
ally changing over a considerable range due to fading. It is undesir-
able that noise or speech on the incoming channel be reradiated on the
outgoing channel. Any tendency for the system to sing must be
avoided. It must be possible to change the amplification looking into
the transmitters over a wide range so as to get a fully modulated out-
put from them, irrespective of the length of the connected lines or
the volumes of the talkers' voices. Furthermore, in some cases, as
where the same radio-frequency band is used for transmission in the
two directions, the radio transmitter tends to interfere with the re-
ceiver at the same end.
A solution of these conflicting requirements necessitates that only
one of the radio paths be connected to the wire network at a time.
This fundamental principle at one stroke wipes out singing, reradia-
tion or echoes, and permits independent adjustments of amplification
in the two radio paths. To apply it, it becomes necessary to employ
voice-current-operated switching devices which connect alternately
the sending or receiving radio channel to the wire line as the sub-
scriber talks or listens, automatically following the conversation and
serving the needs of the subscriber without his volition.
Various mechanisms for carrying out this function have been de-
vised. Some employ mechanical relays for switching while others
use vacuum tubes, but in principle they are much alike. The broader
ideas involved are illustrated in Fig. 6. When the circuit is quiescent,
i.e., neither subscriber speaking, the receiving radio channel is con-
252
BELL SYSTEM TECHNICAL JOURNAL
nected and the transmitter disconnected. Speech coming from the
wire line connects the transmitter and disconnects the receiver. The
positive switching action is, therefore, dependent upon the impulses
of speech from the land line. This arrangement is preferred to the
reverse one of depending upon impulses of speech receiver over the
radio channel. This is because the system must operate on speech
only and not noise, and the speech-to-noise-ratio is usually higher and
more dependable on the wire line than on the incoming radio channel.
This single function of switching-over in response to the subscriber's
voice is the principal and basic function of such devices. There are,
however, many auxiliary features incorporated to guard against false
TERMINAL office:
RADIO
~ PECFIVER
J
J
TRAMS-
WITTER
Fig. 6 — Circuit diagram illustrating operation of voice-operated switching device,
Note: Voice currents coming from the line, rectified in the transmitting detector,
clear the transmitting path by removing short circuit at 55 and short-circuiting
receiving path at TES. Switch at RES is operated by received radio speech or
noise to prevent echoes in the wire lines from reaching transmitting detector.
operation by noise currents and speech current echoes which greatly
increase the ability of the arrangement to operate satisfactorily under
conditions of severe noise or weak speech. These have been described
elsewhere * more completely than would be appropriate in this dis-
cussion.
Viewed from the radio standpoint these voice-operated devices are
of great importance since they permit radio links to be used as trunks
in wire networks without their having to meet the requirements which
wire line trunks must meet. At the present stage of development it
would be practically impossible to provide radio circuits meeting wire
line standards.
* See bibliography 7.
OVERSEAS RADIO EXTENSION 253
Transmission Results
We now come to a consideration of these transoceanic links which
is perhaps the most important one from the standpoint of the service
given and of the engineering development required. It is that of the
general transmission effectiveness and of the continuity of service
which is given. So far as the radiotelephone circuits operating out
of the United States are concerned, this phase of the subject is pretty
well summarized by the charts given in Fig. 7. These show from top
to bottom the continuity of tivo-ivay transmission which has been
obtained over the past year, (1) on the long-wave transatlantic circuit,
(2) on one of the short-wave transatlantic circuits, and (3) on the
short-wave circuit which operates with Buenos Aires. The last named
circuit has been in operation only since the spring of this year.
The black areas show in each case the hours of the day during which
the circuit was commercially usable. The white gaps indicate periods
during which no operation was attempted and for which there are no
data. The dotted-in lines show the periods during which the circuit
was found to be commercially unusable, i.e., the lost time periods.
The following points are to be noted:
1. The long-wave circuit, shown at the top, is poorest during the
summer months. This is because of atmospheric disturbances
due to lightning. Throughout the year shown, the long-wave
circuit was available for service about 80 per cent of the time
2. The North Atlantic short-wave circuit, center figure, was fairly
good last summer but suffered much lost time during the spring
months of 1930. The poor behavior during the spring is ap-
parently due to unusually high solar activity. Such related
phenomena as aurora disturbances in the earth's magnetic field,
and earth currents have been affected similarly. For the year
shown this short-wave circuit was commercially available about
64 per cent of the operating time. Similar experience was had
on the other two transatlantic short-wave telephone circuits,
one of which was operated over a longer period of the day than
that shown.
3. The combination of the North Atlantic of the long-wave and
short-wave circuits gives a much improved result as compared
with either one alone. As is indicated in the diagrams, last
summer when the long wave circuits suffered from "static,"
the short-wave transmission was fairly good; conversely, this
last winter and spring when the short-wave transmission suf-
fered severely from magnetic storm effects, the long-wave cir-
cuit was the mainstay of the service.
254
BE 1. 1. SYS'I'F.M TECH N I CM. JOCKXAL
4. The short-wa\c transmission between New York and Buenos
Aires, as depicted by the bottom chart of Fig. 3, will be seen
LONG WAVE NEW YORK - LONDON STAT ION S W NL i GBT
SHORT WAVE NEW YORK - LONDON STATIONS WMI8.GBU
SHORT WAVE NEW YORK - BUENOS AIRES STATIONS WLO& LSN
,0
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ieJUmij
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2 e
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DEC. I JAN res. MARCH APRIL MAY JUNE
' 1930
COMMERCIAL
Fig. 7 — Chart showing transmission results on long waves — transatlantic; short
waves — transatlantic; short waxes — -South America.
to be more reliable than short-wave transmission across the
North Atlantic. The single short-wave circuit between New
York and Buenos Aires has, since the initiation of this service
OVERSEAS RADIO EXTENSION 255
last spriiiii, been coniniercialK' usable about 97 per cent of the
operating time.
The difference in short-wave transmission east and west across the
North Atlantic and that across the tropical zone, shown in Pig. 7,
is quite in keeping with the general experience of other operating
agencies and is already a well recognized fact in short-wave trans-
mission. There is obviously a radical difference in the character of
the transmission paths involved which requires further survey and
analysis.
Typical Magnetic Storm Effect
It will be noted from the second diagram of Fig. 7 that the inter-
ruption of short-wave transmission across the North Atlantic some-
times continues for several days at a time. These periods have been
found to correspond to disturbances in the magnetic state of the earth
and to be accompanied by the appearance of relatively large differ-
ences of electric potential along the earth's surface. Measurements
which have been carried out on the strength of electric field received
across the Atlantic during such periods and simultaneous records
which have been made of earth potentials shed some light on what
happens during these periods.
There is shown in Fig. 8 observations which were made during a
major effect of this kind which occurred in July, 1928. Short-wave
transmission conditions appeared to have been normal both before
and after the occurrence of this effect. The measurements were made
at New Southgate, England, upon station WND, one of the radio
transmitters at Deal, N. J., used before the present transmitting plant
at Lawrenceville was built. The measurements were made on 18,340
kc. during the normal hours of daylight operation. The upper curve
of the figure shows the variation in received field strength averaged
over the daylight hours for each of the several days shown. Below
the field strength curve there is plotted a record which was made
during this same period of the earth potentials in the vicinity of New
York. This is a smoothed transcript of a record taken on a continu-
ously operated recorder connected in a grounded wire circuit which
extended from New York westward to Reading, Pa., about 100 miles
distant.
It will be observed that the time of minimum field strength coin-
cided approximately with the time of maximum earth potential (the
small wiggles of earth potential are to be neglected since they are due
to disturbances set up by man-made electrical systems). The drop
in the strength of the received field will be observed to be large, of
256
BELL SYSTEM TECHNICAL JOURNAL
the order 35 db. Tlie effect upon transmission lasted several days,
the recovery appearing to have been slower than the initial effect.
A high degree of coincidence has been found to exist between these
adverse effects in short-wave transmission on the one hand, and on
the other hand the appearance of earth currents and abnormalities in
the earth's magnetic field. This is a subject which cannot be ade-
quately treated in the present paper and it is hoped that a report upon
it can be made to the Institute during the forthcoming winter. As is
explained below radio transmission is believed to be largely dependent
fi
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Fig. 8
— Magnetic storm effects, showing drop in field strength and appearance ol
earth potentials.
on the state of ionization of the earth's atmosphere. Earth potentials
are probably also affected by variation in this ionic state. Therefore,
we have in such a recorder a useful check on the transmitting medium
when transmission difficulties are encountered. Such earth potential
observations may prove to be useful in exploring these conditions
more generally throughout the world.
In Fig. 8 each point of the radio data was obtained by averaging
the field strength of the carrier throughout a 24-hour period. Fig. 9,
on the other hand, presents in a more detailed manner the way in
which the field strength varied throughout each of seven days, be-
tween June 24 and July 1, 1930, on transmission from England to the
OVERSEAS RADIO EXTENSION
257
U. S. A. Within this period, there was a magnetic storm. No data
were obtained on June 29. The original curves were obtained with
an automatic recorder, receiving from station GBU of the British
General Post Office during regular operation. In redrafting for pub-
ft«
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6-26-30
Fig. 9 — -Magnetic storm effect, oscillograms of received carrier.
lication, the rapid variations which are characteristic of fading have
been eliminated and only the slow drifts are shown. It will be seen
that the effect of the storm became evident on June 26, the average
signal being 15 to 20 db lower than the preceding day. This condition
continued on the 27th and 28th, on the 30th the signal averaged a little
higher, and on July 1 a recovery had set in. The incompleteness of
25cS Bl'lLL SYSTEM TECHNICAL JOURNAL
the record on three days is caused by the transmitting station shifting
to a different frequency in an attempt to improve conditions. As to
commercial transmission results over this channel during this period:
the first two days were fair, the third day poor, the 27th, the 28th,
29th, and 3()th very poor, and July 1 still rather poor.
The Problem ok the Transmitting Medium
These adverse effects in short-wave transmission are ascribed to
the nature of the medium through which the propagation of the waves
takes place. The short-wave signals which reach a distant point are
carried by waves which have traveled in the upper regions of the
atmosphere, where a condition of ionization exists which causes the
waves to move in a curved path and, finally, to arrive again at the
earth's surface. The ionization in the upper part of the atmosphere
varies with atmospheric conditions and hence its action on the waves
which are passing through it varies from day to night, from season to
season, in a more or less regular manner, on which are superposed
fortuitous variations due to other conditions. The conditions in the
upper atmosphere may be such that two or more waves arrive at a
distant point from the same source after having traversed different
paths. If the length of one of the paths is changing, the resulting
signal from the two waves will pass through a series of maxima and
minima in time, which process is known as fading. This complicated
path condition is present at practically all times, since it is only on
very rare occasions that short-wave signals do not fade in and out.
Furthermore, there appear to be different kinds of fading corresponding
to different transmission paths. For example, the fading on the North
Atlantic short-wave circuits is of a deep slow variety as compared
with the faster and more choppy type of fading experienced on the
north and south circuit between New York and Buenos Aires.
To some extent this fading can be overcome by means of auto-
matic gain control in the radio receiver which causes a steady signal
to be delivered to the listener. However, this does not correct for
the distortion which may be produced by interference between two
transmission components. This distortion may result from a selective
fading of the various frequencies in the voice band and an oscillo-
gram showing this condition is given in Fig. 10 which is taken from
a paper by R. K. Potter.^ These are records of transmission across
the North Atlantic of the voice band occupied by 10 suitably spaced
tones of equal amplitude at the transmitting end. There is shown in
the vertical columns a succession of snapshots which are separated
â– ' See bibliography 19.
OVERSEAS RADIO EXTENSION 250
by intervals of about one-twelfth of a second. By following these
columns down, the progressive change which occurs in the distortion
of the voice band may readily be seen. The worst distortion occurs at
^\
X.
Fig. 10 — Distortion of voice band in short-wave transmission.
times when the carrier itself is blotted out. Tests have indicated that
the use of single side band is of value in minimizing this type of dis-
tortion. Experiments have been in progress for some time looking
to the evaluation of gain to be expected along these lines from the
260 BELL SYSTEM TECHNICAL JOURNAL
introduction of a single side-band system and toward the develop-
ment of single side-band equipment for use at these frequencies.
Another method which might be employed to reduce this type of
distortion is to pick up the signal on a number of antennas spaced
more than about 10 wavelengths apart, since it is found that at points
this far distant from each other, while the general average signal
values are the same, the instantaneous values of the signals are ap-
parently random within the fading limits. By an automatic arrange-
ment for selecting the best signal from, let us say, three antennas
arranged in this manner, voice distortion can be diminished.
During periods of magnetic storms, however, the signals are so
very much reduced in intensity at times that they cannot be heard
above the noise level. There appears to be nothing in the present art
which will fully cope with this situation. Of course, some of the time
which is now lost during these periods may be expected to be regained
by further transmission improvements. As was indicated earlier in
the paper, it is an interesting but rather discomforting fact that these
particularly severe conditions are due to some peculiarity in the con-
dition of ionization as indicated by the magnetic and earth current
disturbances referred to above and by the fact that aurora displays
are likely to be pronounced at these times. Furthermore, it appears
that the transmission is most adversely affected during these times
along paths which pass near the aurora zones surrounding the mag-
netic poles. This is indicated by the marked effect which these storms
had on the North Atlantic circuits while showing only a slight effect
on the South American circuit.
The advantages to be gained by the use of directive antenna sys-
tems were touched on a little earlier in this paper. So far, most of the
gain has been obtained by sharpening the transmission in the hori-
zontal plane. This can be done advantageously only up to a certain
point, corresponding to an antenna spread of from six to ten wave-
lengths—at any rate for transatlantic signals — and representing a
gain when a reflector is used of about 15 db. A further gain of 3 to 5
db can be obtained by sharpening in the vertical plane; and while
a still greater gain can at times be obtained in this manner, the pro-
cedure has so far appeared not worth its possibilities of trouble. This
is due to the fact that with varying conditions in the upper atmosphere,
the waves as they reach the receiving station apparently approach
from different vertical angles and care must be taken not to build an
antenna with such a sharp vertical characteristic that the received
waves will fall on the antenna at such an angle that its calculated
gain cannot be realized. We have, in fact, constructed several an-
OVERSEAS RADIO EXTENSION 261
tennas sharp in the vertical plane, which have given as much as 16 to
20 db gain over a one half wave vertical antenna on local test but
which have given for a signal from a distant point all variations of
gain from this same value down to a loss of 2 db.
Planning the International Use of Frequencies
The problems of the transmitting medium discussed above are
those which have been under study in connection with telephone
transmission across the North Atlantic and between North and South
America. Doubtless further observation and the exploration of other
portions of the earth's surface will disclose a much more complete
picture than it is now possible to present. It is important that fur-
ther data be gathered not alone for the purpose of improving the
transmission results obtained but also for use in agreeing interna-
tionally upon the most effective use of the frequency spectrum for
different services in the interest of the world as a whole.
Of fundamental importance is the question of the frequencies which
are best suited to different distances of transmission. The curves of
Fig. 11 ^ give this relationship between frequency and distance in so
far as it has been disclosed by measurements carried on between North
America and Europe and South America, and also between the Ameri-
can continent and ships plying the Atlantic Ocean. In the construc-
tion of these curves use has been made also of data obtained by other
agencies such as the Radio Corporation and the United States Navy
Department. The curves are reproduced here merely for such use
as they may be in connection with this problem of planning and with
the hope of stimulating the contribution of corresponding data for
other regions of the earth. It should be realized that actually each
curve is the center of a considerable band of frequencies and that
these bands merge one with another.
While experience has indicated that during the adverse transmission
conditions which accompany a magnetic storm some improvement
in transmission can at times be obtained by shifting the frequency.
In general, these effects are found to extend over the entire high-
frequency range now in general use, and shifting frequency does not
dodge them.
In view of the extent to which transoceanic radio links, telegraph
as well as telephone, are dependent upon the use of the higher fre-
quencies, and of the importance of communications to the world as
a whole, it is highly desirable that they be conserved for these longer
distance uses. This has already gained recognition and the 1929
'• See bibliography 12.
262
HI-:LL SYSTh.M TECHNICAL JOCRXAL
Hague Conference of the C.C.I.R. has recommended it as a principle.
The carrying of it out in practice means that, in general, communica-
tions over the shorter distances should be carried out on the lower of
the high frequencies (and possibly at the extreme high frequencies).
It logically calls, also, for making the maximum use of existing wire
networks for overland services, in order to free the radio channels for
uses for which they are most needed. Finally, there is, of course, the
need for coordinating the transoceanic links among themselves and
minimizing unnecessary duplication.
In the Washington, 1927, Convention the world took a construc-
tive step forward in organizing the use of radio channels by blocking
out the high-frequency spectrum in respect to classes of service, thus:
aooo
DISTANCE -STATUTE MILES
Fig. 11 — ^Frequency-distance characteristic.
point-to-point, relay broadcast, mobile services. It is of interest to
note that there is a further line of distinction which might be availed
of for the purpose of reducing interference. As matters now stand,
powerful and expensive stations which can well afford to live up to
the highest standard of frequency stability, radio receiver selectivity,
etc., are intermi.\ed in the frequency spectrum with stations which
cannot justify living up to these standards. Wide differences, in the
caliber of station in accordance with the different needs is, of course,
to be expected. This would appear to call for some grouping of sta-
tions in the various frequency bands in accordance with the frequency
tolerance which they are prepared to meet. Some indication of the
prevalence of interference on these short waves is given by the experi-
ence which has been had in operating the transatlantic short-wave
telephone circuits during the lirst six months of 1930. Of some 3,000
operating hours in which the short-wave circuits were commercially
OVERSEAS RADIO EXTEXSION 263
useful, 110 hours, or about 3 i:)er cent of the time, were lost due to
interference from other stations. The frecjuencies of the interfering
stations were found to differ from their registered frec}uencies by vary-
ing amounts up to hundreds of kilocycles.
The Hague 1929 Conference of the C.C.I.R. recommended that
the frequencies of fixed stations operating in the 6,000 to 23,000-kc.
range be held to 0.05 per cent tolerance and improved to 0.01 per
cent as soon as possible. That this is not an unreasonable recjuire-
ment for large stations is indicated by the following results of meas-
urements made on the four short-wave telephone transmitters at
Lawrenceville, N. J., during the periods of regular operation for the
first half of 1930. Of 2826 measurements of the frequencies of these
transmitters which were made at a measuring bureau 99.75 per cent
were within the ± 0.05 per cent deviation, and 89.1 per cent were
within the ± 0.01 per cent.
The existence of the problems of the transmitting medium and of
the reduction of interference is a reminder of the need which exists
for further quantitative studies of radio transmission throughout the
world and of radio station performance, in the interest of the more
effective use of the radio channels of the world.
Bibliography
1 Ralph Bown, "Some recent measurements of transatlantic radio transmission,"
Proc. Nat. Acad, of Sci., 9, Xo. 7, 221-225; July, 1923.
2. H. D. Arnold and Lloyd Espenschied, "Transatlantic radio telephony," Jour.
A. I. E. E., August, 1923; Bell Sys. Tech. Jour., October, 1923.
3 H. \V. Nichols, "Transoceanic wireless telephony," Electrical Commumcation,
2, Xo. 1, July, 1923.
4. A. A. Oswald and J. C. Shelleng, "Power amplifiers in transatlantic radio teleph-
ony," Proc. L R. E., 13, 313-363, June, 1925.
5 R A. Heising, "Production of single side-band for transatlantic radio telephony,"
Proc. I. R. E., 13, 291-313, June, 1925.
6. Lloyd Espenschied, C. X. Anderson, and Austin Bailey, "Transatlantic radio-
telephone transmission," Bell Sys. Tech. Jour., July, 1925. (Also I. R. E.)
7. S. B. Wright and H. C. Silent, "The Xew York-London telephone circuit,"
Bell Sys. Tech. Jour., 6, 736-749, October, 1927.
8. Frank B. Jewett, "Transatlantic telephony," Scientific Monthly, 25, 170-181,
August, 1927.
9. Ralph Bown, "Transatlantic radiotelephony," Bell Sys. Tech. Jour., 6, 248-
257, April, 1927.
10. K. \V. Waterson, "Transatlantic telephony — service and operatmg features,"
Jour. A. I. E. E., 47, 270-273, April, 1928; Bell Sys. Tech. Jour., 7, 187-194,
April, 1928.
11. O. B. Blackwell, "Transatlantic telephony— the technical problem," Jour.
A. I. E. E., 47, 369-373, May, 1928; Bell Sys. Tech. Jour., 7, pp. 168-186,
April, 1928.
12. Frank B. Jewett, "Some research problems m transoceanic telephony," Proc.
Amer. Soc.for Testing Materials, 28, Part 11, 7-22, 1928.
13. Austin Bailey, S. \V. Uean, and W. T. Wintringham, "Receiving system_^for
long-wave transatlantic radiotelephony," Proc. I. R. E., 16, 1645-1705,
December, 1928. . . „
14. Clifford X. Anderson, "Transatlantic radio transmission and solar activity,
Proc. I. R. E., 16, 297-347, March, 1928.
264 BELL SYSTEM TECHNICAL JOURNAL
15. Clifford N. Anderson, "Solar disturbances and transatlantic radio transmission,'
Proc. L R. E., 17, 1528-1535, September, 1929.
16 S. W. Dean, "Weather phenomena and directional observations of atmos-
pherics," Proc. L R. E., 18, 1185-1192, July, 1929.
17. R. A. Heising, J. C. Schelleng, and G. C. Southworth, "Some measurements
of short-wave transmission," Proc. 1. R. E.., 613, 649, October, 1926.
18. J. C. Schelleng, "Some problems in short-wave telephone transmission," Proc.
I. R. E., 18, 913, June, 1930.
19 R. K. Potter, "Transmission characteristics of a short-wave telephone circuit,"
Proc. I. R. E., 18, 581, April, 1930.
20. H. W. Nichols and Lloyd Espenschied, "Radio extension of the telephone
system to ships at sea," Proc. I. R. E., 193-243, June, 1923.
21. A. E. Harper, "Directional distribution of static," Proc. I. R. E., 17, 1214-
1225, July, 1929.
22. W. Wilson and L. Espenschied, "Radiotelephone service to ships at sea," Jour.
A. I.E. E., 49, 542, July, 1930.
23. T. G. Miller, "Transoceanic telephone service — general aspects," Jo7tr. A. I.
E. E., 49, 107, February, 1930.
24 Ralph Bown, "Transoceanic telephone service — -short-wave transmission," Jour.
A. I.E. E., 49, 385, May, 1930.
25. A. A. Oswald, "Transoceanic telephone service — -short-wave equipment," Jour.
^. 7. £. £., 49, 267, April, 1930.
26. F. A. Cowan, "Transoceanic telephone service — short-wave stations," Jour.
A. I. E. E. (forthcoming).
27. P. Craemer, "Der Weltfernsprechverkehr," E. T. Z., 50, 959-963, July, 1929.
28. P. Craemer, "The geographical implications of the world telephone network,"
Paper No. 206, World Engineering, Congress, Tokio, 1929.
29. E. H. Shaughnessy, "Rugby radio station," Jour. P. 0. E. E., 19, 373-382,
January, 1927; El. Rev., 98, 753-756, May 7, 14, 21, 1926; Elect., 96, 468-469,
April 23, 1926.
30. A. G. Lee, "Transatlantic telephony," Jour. Tel. & Tel., 13, 92-93, February,
1927; Jour. P. 0. E. E., 19, 74-75, April, 1926; Jour. Tel. & Tel, 12, 150-151,
April, 1926.
31. R. V. Hansford, "London-New York telephone circuit," Jour. P. 0. E. E., 20,
55-64, April, 1927.
32. T. F. Purves, "Ship and shore telephony," Electrician, 104, 516-517, April 25,
1930.
33. T. F. Purves, "Ship-shore radiotelephony," El. Rev. (London), 106, 865-866,
929-930, May 9-16, 1930.
34. A. S. Angwin, "Ship-and-shore terminal equipment," Electrical Communication,
9, 56-61, July, 1930.
35. T. F. Purves, "Inaugural address," Jour. I. E. E., 68, No. 396, pp. 1-16, Decem-
ber, 1929.
Some Optical Features in Two-Way Television *
By HERBERT E. IVES
A comprehensive description of the two-way television system now being
demonstrated between the American Telephone and Telegraph Company
building, and the Bell Telephone Laboratories, in New York City, has been
published elsewhere.^ Part of that account gives the essential features of the
optical arrangements whereby the users of the apparatus are appropriately
lighted, and are assured against visual discomfort from the scanning opera-
tion. Since the apparatus was first installed, however, some important
changes have been made in the distinctively optical features, whereby the
performance of the system has been notably improved, and its operation
considerably simplified. These changes deserve description, and the pres-
ent account is mainly concerned with them, although for the sake of com-
pleteness some details previously described are included.
IT IS an inherent feature of the two-way television system that either
user is continuously scanned as he views the image from the distant
station. The beam scanning method,^ by which a beam of light sweeps
over the subject's face, enables the scanning operation to be performed
with a minimum amount of light. Even so, because of the relatively
low intensity of the television image, it is necessary to reduce the in-
tensity of the scanning beam in every way possible. In the two-way
apparatus as first operated, advantage was taken of the fact that the
photoelectric cells employed, which were of potassium, were principally
sensitive to blue light. The scanning beam derived from a high power
arc lamp was accordingly passed through a deep blue filter, which
reduced the photoelectric efficiency of the beam very little, but because
of the relatively low visual value of blue light, effectively reduced the
brightness of the beam many times. The user of the apparatus saw,
above the incoming image, merely a mild blue spot of light, which did
not interfere with his vision.
A disadvantage of the use of blue light, which was anticipated, and
found in practice to be quite real, was that dark, tanned, or ruddy
complexions were rendered as altogether too dark, in comparison with
whites such as the ordinary linen collar. The effect is precisely that
encountered in the earlier photographic processes before color sensitive
plates and color filters were available. While this defect was mini-
mized by the use of a dark background, and to some extent by chopping
off the highlights by electrical means, it was recognized as undesirable.
* Jour. Optical Soc, Feb. 1931.
^Bell Sys. Tech. Jour., July 1930.
2 Jour. Optical Sac, March 1928.
265
266 BULL SYSTEM TECHNICAL JOURNAL
One recent improvement in the apparatus is a change in the nature of
the scanning Hght, whereby, without sacrificing the general principle of
using visually inefficient hut photoelectrically efficient radiation, the
proper balance of tone values in the face is restored. This has been
accomplished by adding to the battery of blue sensitive potassium cells,
a group of red sensitive caesium oxide cells, and scanning by purple
instead of blue light, that is, both ends of the visible spectrum are
used in place of one end.
In making this change, a number of others were involved, most of
which resulted in simplification or improvement. One important
alteration was the substitution for the arc lamps previously employed,
of incandescent lamps of a type available from motion picture projec-
tion practice, as shown in Fig. 1. The lamp employed has for its radia-
tor, four vertical helical coils of tungsten wire, and is furnished with a
reflector which images the coils back on the intervening spaces. An
efficient condenser system throws a brilliant rectangular image on the
back of the scanning disc, which is substantially uniform over the
whole field. With this unit, the scanning beam as it leaves the pro-
jection lens is somewhat larger in diameter than the beam as produced
from the arc. Consequently, for positions away from the focused
image of the disc holes, the scanning beam is larger than before, with
some resultant loss in the range of sharpest definition. Since, however,
the user of the two-way apparatus is seated in a fixed chair, he has little
opportunity to move far out of the plane in which the disc holes are
focused, so that this objection is not serious. The advantages of this
substitution were two-fold. First was a great gain in simplicity of
operation and maintenance. Second, the incandescent lamp, being a
lower temperature radiator, radiates relatively many times as much
red light as does the arc, for the same amount of blue. Consequently,
once an incandescent lamp unit was found which gave the amount of
blue light required for the potassium cells, the great e.xcess of red light
made possible the use of relatively few Ccesium oxide cells. Since
these are intrinsically somewhat more sensitive than the potassium cells,
the net result was that a red signal comparable with the blue signal
could be added by the installation of only two caesium cells, each of less
than half the electrode area of the potassium cells.
It was found most convenient to mount the two Cccsium oxide cells
directly in front of the observer, to either side of the microphone, and
above the opening in the booth through which the scanning beam
enters, and through which the incoming image is seen. This arrange-
ment is shown in Fig. 2. The only objection to placing the cells in this
position is that they encroach somewhat into the region where reflec-
SOME OPTICAL FEATURES IN TWO-WAY TELEVISION 267
lions of the cells (which are virtual light sources) are likely to be seen
reflected in eyeglasses. Since, however, the head is normally directed
somewhat downward, cells placed in these upper corner spaces are not
serious offenders in this respect.
Two other features of the two-way system which needed revision
when the caesium cells were adopted, were the variable angle prisms
used to direct the scanning beam upward or downward, depending on
F~ig. 1 — Incandescent lamp used for scanning light.
the user's height, and the general illumination of the television-tele-
phone booth. As to the variable angle prisms, the only change called
for was the substitution of achromatic prisms, corrected for deep red
and blue light, in order to prevent the scanning beam from breaking
effectively into two beams for large angles of deviation. The problem
of general illumination of the booth is principally the choice of a color
of light which shall affect neither the potassium nor the ca'sium cells.
For this purpose, a monochromatic yellow-green was chosen, secured
by covering all the lights with a combination of orange and signal green
glasses. The potassium cells are insensitive to this color of light, and
268
BELL SYSTEM TECHNICAL JOURNAL
the caesium cells were rendered so by placing over them, windows
covered with a deep purple gelatin. This choice of illumination color
made possible a satisfactory general level of illumination of the booth
and the surroundings of the image without introducing spurious signals.
The transmissions of the purple filters, the response curves of the
potassium and caesium oxide cells, the radiation curve of the incandes-
cent lamps used for the scanning beam, and the transmission curves of
the glasses used over the lamps for general illumination, are shown
Fig. 2-
-Interior of two-way television booth showing location of two caisium cells
above and to either side of scanning and viewing aperture.
in Fig. 3. Comparing these with the response curve of the eye, also
shown in the same figure, it will be evident how the general problem of
securing photoelectric signals of maximum efficiency without interfering
with the general quality of the image, or desirable conditions of illum-
ination, has been secured.
Before going on to describe some of the optical features at the re-
ceiving end, we may pause to discuss the improvements in the tele-
vision signal which have been introduced by the changes just described.
There is, of course, a substantial gain in the steadiness of the image due
SOME OPTICAL FEATURES IN TWO-WAY TELEVISION
269
100
80
60
40
20
100
80
I-
5 60
O
UJ
a. 20
(
â– \
\
EYE
—
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>
1
V
y
s
^
j^
s
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\
4 MM SIGNAL
PURPLE
/
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WITH FILTER
-fPURPLE gelatin)
s
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BLACK BODY RADIATION
jTUNGSTEN LAMP 2848° K)
y
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8
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GREEN + ORANGE
GLASS
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4,000 5,000 6,000 7,000 8,000 9,000 10,000 11,000
\ IN ANGSTROMS
Fig. 3— Spectral characteristics of the scanning, viewing and illumination elements
in two-way television system.
270 BELL SYSTEM TECHNICAL JOURNAL
to the elimination of the arc lamps, much of whose effective radiation
was from the arc stream which always wanders somewhat. The
chief gain, however, is in the tone quality of the image of the face.
The difference is very clearly shown if shutters are arranged so that
either the potassium or the ca'sium cells may be used alone, alternately,
and can then be quickly exposed together. With the potassium cells
alone, as already noted, flesh tints are in general too dark, and tanned
or ruddy complexions show unnatural contrast with the whites. High-
lights due to reflection on the skin are often observed to be out of scale,
with a resultant effect of mottling of the skin. With the ceesium cells
alone, on the other hand, the flesh tints are in general too light, and
faces are apt to appear very flat. These differences were anticipated,
but others not so obviously to be expected, have been observed. For
instance, with the caesium cells, the pupil and iris of the eye are brought
out with rather startling blackness, while with the potassium cells.
the detail around the eyes is apt to be lost. The most satisfactory re-
sults are obtained with both sets of cells acting, for, as was hoped, the
combination of the two ends of the spectrum, gives, in the case of the
face, an effect very like that which light from the middle of the visible
spectrum would give, that is, an '"orthochromatic" image, as it would
be described in photography, while the definition of important points,
such as the eyes, is distinctly improved.
Passing now to the receiving end of the two-way television apparatus
we recall that in the apparatus as originally set up and described, a
simple disc with a spiral of holes was used, immediately behind which
was a neon lamp with a large flat water-cooled electrode. On con-
tinued operation, it was found that the heavy current demanded in
these lamps, in order to secure an image of sufficient brightness, caused
rapid sputtering on the closely adjacent glass wall, necessitating fre-
quent renewals of lamps. A very radical change in the disc and lamp
design has been made by which this undesirable situation has been
remedied.
The change in the disc consists in substituting for the simple Nipkow
disc, with its spiral of holes, an alternative form, suggested also by
Nipkow, in which each disc hole has associated with it a condensing
lens, positioned so as to focus, in combination with a fixed collimating
lens, and image of the source on the disc hole. The optical arrange-
ment is shown in Fig. 4, and a photograph of the disc with lenses and
lamp in place in Fig. 5. Referring to Fig. 4, D represents in section
the simple disc with a spiral of holes, h ; / represents a small short focus
lens, fixed in position with respect to // at a distance equal to its focal
length; L represents a fixed lens of diameter large enough to cover the
SO.\[E OPTICAL FEATURES IN TWO-WAV TF.LEVISIOX 271
Fig. 4 — Sertion of disc with lens system for utilizing small area light source.
Fig. 5 — Disc with condensing lenses as used at the receiving ends of two-way tele-
vision system.
272 BELL SYSTEM TECHNICAL JOURNAL
entire frame of the picture and the lenses /; P represents the glow lamp
electrode. A great advantage of this optical arrangement is that the
cathode of the glow lamp can be made quite small, and can be removed,
as shown, to a considerable distance from the glass wall of the contain-
ing tube. In consequence of these changes in lamp design, a very high
current density can be obtained for a relatively low expenditure of
energy, with at the same time a long lamp life.
The condenser lens disc is observed exactly as the simple disc, by
the eye placed at E. According to Nipkow, when lenses are used on
the disc, the holes should be covered with diffusing material. This is
not necessary in the present case, because in the two-way booth, the
observer has very little latitude of motion, and it is only necessary that
his eyes lie in the overlapping cones of rays from the extreme holes in
the field. By making the lenses / of large diameter compared with
their focal length, the solid angle through which an image is visible
is entirely adequate.
The general characteristics of the lamps used are shown in Figs. 4
and 5. The cathode is a heavy slug of copper, into which a hollow
cylindrical aluminium electrode is screwed, shielded from the copper by
mica and glass. Because of the large mass of the copper, the water-
cooling is no longer necessary. With lamps of this type, the amplifier
circuit used before makes it possible to obtain images of much greater
brilliancy, whereby the contrast between the image and the scanning
light is still further increased beyond what w^as before found satisfac-
tory. This margin of brightness is so large that it has been found
possible to use lamps filled with helium in place of neon, giving a much
whiter image, more pleasing to some people.
B ayes' Theorem
An Expository Presentation *
By EDWARD C. MOLINA
B AYES' theorem made its appearance as the ninth proposition in an
essay which occupies pages 370 to 418 of the Philosophical
Transactions, \^ol. 53, for 1763. An introductory letter written by
Richard Price, "Theologian, Statistician, Actuary and Political
Writer," ^ begins thus:
" I now send you an essay which I have found amongst the papers of our deceased
friend Mr. Bayes, and which, in my opinion, has great merit, and well deserves to be
preserved."
A few lines farther on Price says:
"In an introduction which he has writ to this Essay, he says, that his design at
first in thinking on the subject of it was, to find out a method by which we might judge
concerning the probability that an event has to happen, in given circumstances,
upon supposition that we know nothing concerning it but that, under the same
circumstances, it has happened a certain number of times, and failed a certain other
number of times."
"Every judicious person will be sensible that the problem now mentioned is by
no means merely a curious speculation in the doctrine of chances, but necessary to be
solved in order to a sure foundation for all our reasonings concerning past facts, and
what is likely to be hereafter."
No one will dispute the importance ascribed to Bayes' problem by
Price; in fact, a paper by Karl Pearson on an extension of Bayes'
problem is entitled "The Fundamental Problem of Practical Sta-
tistics." Opinions differ, however, as to the validity and significance
of the solution submitted in the essay for the problem in question. In
view of this situation I shall limit myself today to an exposition of the
fundamental characteristics of the problem Bayes' theorem deals with
and shall give no consideration to its interesting applications.
The exposition may be outlined as follows: after specifying the class
of problems to which Bayes' theorem pertains I shall :
* Read before the American Statistical Association during the meeting of the
American Association for the Advancement of Science in Cleveland, Ohio, December,
1930.
1 These titles are associated with the name of Price in the frontispiece portrait of
him bound with the December, 1928, issue of Biometrika.
273
274 BELL SYSTEM TECHNICAL JOURNAL
I. Discuss briefly two problems each of which will emphasize one of
two kinds of a priori probabilities which should be constantly borne in
mind when Bayes' theorem is under consideration,
II. Partially analyze a certain ball-drawing problem which will not
only serve as an introduction to the algebra of Bayes' theorem but will
later help to throw light on its significance,
III. Present Bayes' problem and the related theorem,
IV. Make some remarks on the value of the theorem and the contro-
versies which it raised.
In carrying out this plan I shall find it convenient to ignore the
historic order of events.
When probability is the subject under consideration one anticipates
problems such as: A coin is about to be tossed 15 times; what is the
probability that heads will turn up seven times? A sample of 100
screwdrivers is to be taken from a case containing 1000 screwdrivers of
which 300 are known to be defective; what is the probability that the
sample will contain 25 defectives?
These are direct, or a priori, probability problems. In each of them
the nature of a game, or an experiment, is specified in advance and
then a question is asked relating to one, or more, of the possible out-
comes of the game or experiment. Problems of this type have occupied
the attention of mathematicians since the days of Pascal and Fermat,
the creators of the mathematical theory of probability.
An inverse class of problems of great practical significance, called a
posteriori probability problems, came into prominence with the publi-
cation of Bayes' essay. In these we find specified the result or out-
come of a game which has been played, whereas the question then
asked is whether the game actually played was one or some other of
several possible games. This type of problems is usually stated as
follows :
"An event has happened which nuist have arisen from some one of a given number
of causes: required the probability of the existence of each of the causes."
I
Consider this example: during his sophomore year Tom Smith
played on both the baseball and football varsity teams; we have been
informed that he broke his ankle in one of the games; what are the a
posteriori probabilities in favor of baseball and football, respectively,
as the baneful cause of the accident?
Evidently the answer depends on the number of baseball and
football games played during their respective seasons and also on the
likelihood of a man breaking an ankle in one or the other of these two
games. As a concrete case assume that:
BAYES' THEOREM 275
1. At Smith's college an equal number of baseiiall and football games
are played per season;
2. Statistical records indicate that if a student participates in a base-
ball game the probability is 2/100 that he will break an ankle
and that, likewise, the probability is 7/100 for the same con-
tingency in a football game.
In view of the first of these two assumptions our conclusions as to the
cause of the accident may be based entirely on the information con-
tained in the second assumption. The odds are two to seven, so that
the a posteriori probabilities regarding the two admissible causes are:
For baseball, 2/(2 + 7) = 2/9.
For football, 7/(2 + 7) = 7/9.
Now consider this other example. A lone diner amused himself
between courses by spinning a coin. We elicited from the waiter that
in 15 spins, heads turned up seven times. Moreover, from our point
of observation, the size of the coin indicated that it was either a silver
quarter or a ten-dollar gold piece. What are the a posteriori proba-
bilities in favor of the silver quarter and the gold piece, respectively?
If the lone diner were a professor from one of our eastern universities
we would not hesitate a moment in declaring that the coin spun was a
quarter. But it happens that the gentleman was a member of the
Cleveland Chamber of Commerce, dining at the Bankers' Club. We
must, therefore, give the matter more careful consideration. The
number of quarters and gold pieces usually carried by a banker and the
probabilities of obtaining the observed result by spinning coins are
relevant; let us assume, therefore, that:
1. The small change purse of a Cleveland financier contains, on the
average, ten-dollar gold pieces and quarters in the ratio of
eight to three.
Moreover, we may assume (in fact we know) that:
2. If either a quarter or a gold piece is spun 15 times, the probability
that heads will turn up seven times is approximately 1/5.
The second of these two items of information makes the a posteriori
probabilities depend entirely on the first item. Clearly the odds are
eight to three and we conclude;
For a quarter, a posteriori probability = 3/ {3 + 8) = 3/11.
For a goldpiece, a posteriori probability = 8/(3 + 8) = 8/11.
276 BELL SYSTEM TECHNICAL JOURNAL
Now regarding the general a posteriori prol)lem,
"An event has liai)i)ened which must liave arisen from some one of a given number
of causes: required the probability of the existence of each of the causes,"
what do the two examples we have just considered suggest? In both
problems we inquired into:
1. The frequency with which each of the possible causes is met with
BEFORE THE OBSERVED EVENT HAPPENED. This frequency
is called the a priori existence probability for the corresponding
cause.
2. The probability that a cause, if brought into play, would reproduce
the observed event. This probability will hereafter be referred
to as the a priori productive probability for the cause in question.
In the case of the broken ankle, the a priori existence probabilities were
equal and took no part in our conclusion; we based the a posteriori
probabilities entirely on the a priori productive probabilities. We did
just the opposite with reference to the coin spun by the Cleveland
financier; on account of the equality of the a priori productive proba-
bilities we deduced a posteriori probabilities in terms of the unequal a
priori existence probabilities.
It is apparent that our two examples represent extreme cases. In
general, the solution of an inverse or a posteriori problem, involving a
number of causes, one of which must have brought about a certain
observed event, depends on both sets of direct, or a priori probabilities.
Those of the first set give the frequency with which the various causes
were to be expected before the observed result occurred ; those of the
second set give the frequencies with which the observed result would
follow from the various causes if each were brought into play.
II
Bearing in mind the two distinctly different sets of a priori proba-
bilities required in arriving at a posteriori conclusions regarding the
possible causes of an observed event, we must now give some thought
to the algebra of the subject before taking up Bayes' problem and
theorem. For this purpose consider the following bag problem:
A bag contained M balls of which an unknown number were white.
From this bag N balls were drawn and of these T turned out to be
white. What light does this outcome of the drawings throw on the
unknown ratio of the number of white balls to the total number of
balls, M, in the bag? Let x be this unknown ratio.
Two cases of this problem may be considered:
BAYES' THEOREM 277
Case 1. — After a ball was drawn it was replaced and the bag was
shaken thoroughly before the next drawing was made;
Case 2. — A drawn ball was not replaced before the next drawing.
These two cases become essentially identical when the total number of
balls in the bag is very large compared with the number drawn.
Case 1 will serve as an introduction to Bayes' problem; later we will
find it highly desirable to consider Case 2.
We are confronted with (.1/ + 1) possible hypotheses or causes
before the drawings took place:
1 — the unknown value of x is Xo = 0/M,
2 — the unknown value of x is .ti = 1/M,
3 — the unknown value of x is .T2 = 2/i/,
k -{- I — the unknown value of x is Xh = k/M,
J/ + 1 — the unknown value of x is Xm = M/AI = 1.
Let w{xk) be the a priori existence probability for the ^'th hypothesis;
by this is meant the probability in favor of the ^'th hypothesis based on
whatever information was available regarding the contents of the bag
prior to the execution of the drawings.
Let B{T, N, Xk) be the a priori productive probability for the ^'th
hypothesis ; by this is meant the probability of obtaining the observed
result (r whites in N drawings) when the value of x is klM.
Then, the a posteriori probability, or probability after the observed
event, in favor of the ^'th hypothesis is
P w{xk)B{T, N, Xk) ..x
E w{x,)B{T, N, xu)
t=o
For Case 1 of our bag problem we have
B{T, N, X,) = [t) ■^'''^^ ~ ''"'•^"'"'''
where Ij.) represents the number of combinations of N things
2 This is the Laplacian generalization of Bayes' formula, although in some text-
books it is referred to as " Bayes' Theorem." A relatively short demonstration of it is
given by Poincare in his Calcul des Probahilites. See also Fry, Probability and Its
Engineering Uses, Art. 49.
278
BELL SYSTEM TECHNICAL JOURNAL
taken T at a time. Substituting in (1) we obtain, after canceling
from numerator and denominator the common factor ( 7^ j .
Pk =
(2)
If in eciuation (2) we give k successively the values a, a + 1, a + 2,
• • • h — \, b and add the results we have
Pa + Pa + 1 +
+ Pt
or
P{Xa, Xb)
L w{Xk)Xk''X^ - Xk)-^''
£; w{xk)xk'^{i - Xk)""' '^
A=0
(3)
for the a posteriori probability that the unknown ratio of white to
total balls in the bag lies between a/AI and b/M; both inclusive.
Ill
Bayes' Problem
Consider the table represented by the rectangle A BCD in Fig. 1.
On this table a line OS was drawn parallel to, but at an unknown
distance from, the edges AD and BC. Then a ball was rolled on the
table N times in succession from the edge AD toward the edge BC.
As indicated in the figure, it was noted that T times the ball stopped
rolling to the right of the line OS and .V - T times to the left of that
line.
What light does this information shed on the unknown distance
from ^L> to OS? In more technical terms, what is the a posteriori
probability that the unknown position of the line OS lies between any
two positions in which we may be interested?
C
S
B
1
O
(N-T)
2
.
. .
7^
.
1
■•
U
D
O
A
I'ig. 1.
BA VES' TIIF.ORKM 279
Each rolling of the ball was executed in such a manner that the
probability of the ball coming to rest to the right of OS is given by the
unknown ratio of the distance OA to the length BA of the table;
likewise, the probability of the ball stopping to the left of OS is given
by the ratio of the distance BO to the length BA .
Set X = OA/BA, 1 - x = BO/BA.
The only difference between this problem and the bag of balls
problem is that now the possible values of x are not restricted to the
finite set 0/M, 1/M, 2/M, • • • (M - 1)1 M, M/M; in the table problem
X may have had any value whatever between the limits and 1.
Therefore equation (3) will answer the question asked provided we
substitute definite integrals in place of the finite summations. This
substitution gives us, for the desired a posteriori probability that x had
a value between ;Vi and X2, the formula
I iv(x)x'^{l — x)''^~''^dx
P(xu X,) = -^^ (4)
Jo
Equation (4) is useless until the form of the a priori existence function
w{x) is specified; this depends on the way in which the line OS was
drawn. Bayes assumed that the line OS, of unknown distance from
AD, was drawn through the point of rest corresponding to a preliminary
roll of the ball. This amounts to postulating that all values of x,
between and 1, were a priori equally likely. In other words, with
Bayes, the a priori existence function iv{x) was a constant which,
therefore, did not have to be taken into consideration.^ Thus, instead
of equation (4), Bayes gave the equivalent of the following restricted
formula:
r \r''(l - xY'^dx
P{xu X.) = 7i ; (5)
I .r^(l - xy-'^dx
I say "the equivalent of" (5) because in Bayes' day definite integrals
were expressed in terms of corresponding areas.
Equation (5) constitutes Proposition 9 of the essay, but is usually
referred to as Bayes' theorem.
3 The existence function u^.v) does not appear eitlier exi^licitly or implicitly any-
where in Bayes' essay. This fact raises the question as to whether or not Bayes had
any notion of the general problem of causes.
280 BELL SYSTEM TECHNICAL JOURNAL
IV
Equation (5) is a very beautiful formula; Init we must be cautious.
More than one high authority has insinuated that its beauty is only
skin deep. Speaking of Laplace's generalization and extension of the
theorem, George Chrystal, the English mathematician and actuary,
closed a severe attack on the whole theory of a posteriori probability ^
with the statement that "Practical people like the Actuaries, however
much they may justly respect Laplace, should not air his weaknesses
in their annual examinations. The indiscretions of great men should
be quietly allowed to be forgotten."
Chrystal's advice as to the attitude one should assume toward "the
indiscretions of great men" is excellent, but in the case under con-
sideration, it was the plaintiff rather than the defendant who com-
mitted indiscretions; this is discussed in a paper by E. T. Whittaker *
entitled "On Some Disputed Questions of Probability."
The discussions and disputes, which began shortly after the birth of
the formula in 1763 and which have not as yet subsided, may be
divided into two classes:
\. Discussions concerning problems in which it is known that the a
priori existence function is not a constant.
2. Discussions concerning problems in which nothing whatever is
known concerning the a priori existence function.
The discussions of Class 1 are out of order in so far as Bayes' theorem
is concerned; recourse should be had to formula (4), Laplace's generali-
zation of the Bayes' theorem, when it is known that w{x) is not a
constant. Failure to differentiate explicitly between equations (4)
and (5) has created a great deal of confusion of thought concerning the
probability of causes. The discussions of Class 2 have centered on
what Boole called "the equal distribution of our knowledge, or rather
of our ignorance," that is to say "the assigning to different states of
things of which we know nothing, and upon the very ground that we
know nothing, equal degrees of probability." Regarding the legiti-
macy of this procedure Bayes himself contributed a very important
scholium which appeared in his essay on pages 392 and 393. The
argument in this scholium, based on a corollary to Proposition 8 of the
essay, may be summarized as follows:
Assuming that all values of x are a priori equally likely and that the
N throws of a ball on the table have not yet been made, the probability
■• "Oil Some P'undamental Principles in the Theory of ProbabiHty," Transactions
oj the Actuarial Society of Edinburgh, Vol. 11, No. 13.
^ Transactions of the Faculty of Actuaries in Scotland, \o\. \TII, Session 1919-1920.
BAYES' THEOREM 281
that T times the ball will rest to the right of OS and that the remaining
N — T times it will rest to the left of OS is (as shown in the corollary)
a result in which T does not appear. In other words, any assigned
outcome for the throws is no more, or no less, likely than any other
outcome, if a priori all values of x are equally likely. But, wrote
Bayes in the scholium, when we say that we have no knowledge
whatever a priori regarding the ratio .v, do we not really mean that we
are in the dark as to what will be the outcome when we proceed to
make N throws? If so, then equat-on (6) justifies the assumption that
a priori all values of x are equally likely.
To clinch his argument it must be shown that the converse of
equation (6) is true. That is, it must be shown that, if any outcome
of throws not yet made is as likely as any other, then any value of x is a
priori as likely as any other. This converse theorem was submitted
to Dr. F. H. Murray who obtained an elegant proof based on a theorem
of Stieltjes.^
In view of Bayes' corollary and his scholium, an analysis of our bag
problem with reference to the "equal distribution of our knowledge, or
ignorance" is in order.
Consider again Case 1 where each drawn ball is replaced in the bag
before the next drawling is made.
Assuming each of the {M +1) permissible hypotheses to be a priori
equally likely, the probability that A^ drawings, not yet made, will
result in T white and N — T black balls is
P^t,j^(i)i^yi^-^Y-. (7)
tTo .!/+,! \ T J \M J \ Mj
Equation (7) is not, in general, independent of T^ so that any one
assigned outcome of N drawings is not as likely as any other outcome.
This result is disturbing; at first sight it seems to discredit Bayes'
scholium. \\'e must, therefore, look into the matter more closely.
Bayes' problem corresponds to drawings from a bag containing an
infinite number of balls. Therefore, even if drawm balls are replaced,
^Bulletin of the American Mathematical Society, February 1930.
^ Consider, for example, the case of M = 2. Equation (7) reduces to
a result which is not independent of T.
282 BELL SYSTEM TECHNICAL JOURNAL
the chance of a particular ball being drawn more than once is zero.
But when N drawings with replacements are made from a bag con-
taining a finite number, M, of balls, we are by no means certain of
drawing N different balls; a particular white ball may be drawn several
times over and, likewise, a particular black ball may appear more than
once. It is not surprising, therefore, that Case 1 of the bag problem
does not confirm Bayes' corollary.
Consider now Case 2, where the drawn balls are not returned to the
bag. If k of the total balls are white and the rest black, the probability
that a sample of A^ balls from the bag will contain T white and N — T
black is
k\(M - k\ I / M'
Tj \N- Tj I \ N
Hence, if the permissible values 0, 1, 2, 3, • • • M for k are all equally
likely a priori, we obtain instead of (7),
a result independent of any assigned value for T and identical with the
result in the corollary to Proposition 8 of the essay.
Summary
Bayes' theorem is the answer to a special case of the general problem
of causes. The special case postulates that the a priori existence
probabilities for the various admissible causes of an observed event are
equal.
In the essay Bayes recommends that his theorem be adopted when-
ever we find ourselves confronted with total ignorance as to which one
of several possible causes produced an observed event. To justify this
recommendation Bayes takes the attitude that: a state of total
ignorance regarding the causes of an observed event is equivalent to
the same state of total ignorance as to what the result will be if the
trial or experiment has not yet been made. This interpretation is a
generalization of the fact that in his billiard table problem, the as-
sumption of equal likelihood for all possible positions of the line OS,
gives equal probabilities for the various possible outcomes of a set of N
ball rollings not yet made.
Laplace, Poincare and Edgeworth ^ have shown that the a priori
existence inncXAon iv{x), which appears in the Laplacian generalization
8 Laplace: "Oeuvres," Vol. 9, p. 470. Poincare: "Calcul des Probabilites," 2d
edition, p. 255. Rowley: "F. Y. Edgeworth's Contributions to Mathematical
Statistics," pp. 11 and 12.
BAYKS' THEOREM 283
of Bayes' theorem, is of negligible importance when the numbers N
and T are large. Therefore, when this condition holds, one need not
hesitate to use Ba>es' restricted formula for the solution of a problem
of causes.
The transmission, by Price, of Bayes' posthumous essay to the
Royal Society marked an epoch in the history of the literature on
probability theory. As mentioned at the beginning of this paper,
Karl Pearson has called the extension of Bayes' problem the "Funda-
mental Problem of Practical Statistics."
Extensions to the Theory and Design of
Electric Wave-Filters
By OTTO J. ZOBEL
'l"he problem of terminal wave-filter impedance characteristics is con-
sidered in this paper, in particular that of obtaining an approximately con-
stant wave-filter impedance in the transmitting bands of a wave-filter of any
class, which is of importance where the wave-filter is terminated by a
constant resistance, the usual case. The solution obtained is based upon the
repeated use of the methods of deriving wave-filter structures which gave the
.l/-types, combined with composite wave-filter principles. The results are
wave-filter transducers which at one end have standard "constant k" iniage
impedances and at the other have image impedances which can theoretically
be made constant in the transmitting bands to any degree of approximation
desired. Practical fixed structures are shown.
Parts I and II give this derivation and composition of wave-filter struc-
tures. Two allied subjects, respectively, the designs of networks which
simulate the impedances of wave-filters, and of loaded lines, are dealt with in
Parts III and IV, such designs making use of the previous results.
The four Appendices contain new reactance and wave-filter frequency
theorems, particular fixed transducer designs and certain equivalents; also, a
chart for determining terminal losses at the junction of such a fixed wave-
filter transducer and a resistance termination. This chart siipplements
those previously given in a chart method of calculating wave-filter trans-
mission losses.
o
Introduction
NE important problem which frequently arises in wave-hlter
design is that of obtaining a terminal wave-filter impedance
which is approximately a constant resistance at all frequencies in the
transmitting bands. This ideal impedance characteristic is desirable
where a wave-filter is terminated by such a constant resistance, as is
usually the case. Under these ideal conditions, for frequencies in the
transmitting bands all terminal reflection losses are avoided, and there
are no impedance irregularities at the terminal junction to be reflected
back through the wave-filter and produce objectionable impedance
irregularities at the other end.
The design of ladder type wave-filters of any class, ^ regarded from
either the theoretical or the practical standpoint, involves taking into
consideration two standard image impedances; and the internal or
main part of a composite wave-filter structure, called the mid-part,
usually has the equivalent of one or the other of these image impedances
at each terminal. These two standard image impedances are the image
^"Theory and Design of Uniform and Comi)usile IClcclric W a\ c-Filters," (J. J.
Zobcl, B. S. T. J., January, 1923.
2S4
ELECT lU C WA\ '!â– . - EIL TERS 285
impedances- at the two mid-points, mid-series and mid-shunt, of the
"constant k'' wave-filter of that class. As defined in the first paper
referred to, a "constant k'' wave-filter is a reactance network of ladder
type, the product of whose series and shunt impedances is k"^ = R^, a
constant independent of frequency, where k has the significance of
being the impedance of the corresponding uniform line. It is well
known that these standard, or "constant k," image impedances vary
greatly with frequency over all the transmitting bands and are therefore
far from satisfactory as terminal wave-filter impedances. What is
needed at a terminal having such an image impedance is a terminal
wave-filter transducer of the same class w^hich at one end can be
joined without impedance irregularity to the standard termination and
which at the other end has a desirable terminal image impedance.
Actually, this amounts to terminating a composite wave-filter in a
section which has at the final terminals the image impedance desired.
We may set up the ideal for this purpose as follows :
The ideal terminal wave-filter transducer of any class is a dissymmetrical
wave-filter network having at one end an image impedance equal at all
frequencies to the standard mid-series or mid-shunt image impedance of
the ''constant ^" wave-filter and at the other end an image impedance
which has approximately the same constant resistance value (k = R) at all
frequencies in the transmitting bands.
^^'hile the principal function of such a transducer is to furnish the
desired terminal image impedance, its wave-filter propagation charac-
teristics would also be useful.
The first approximate solution previously obtained was by means of
,l/-type wave-filter terminations;'^ that is, the terminal transducer in
this case w^as a single mid-half section of an M-type wave-filter whose
parameter m is in the neighborhood of m = .6. Such a section has at
one end one of the two standard image impedances referred to above
for all frequencies. At the other end its image impedance has the same
constant resistance value within about 4 per cent over 86 per cent of
every transmitting band and this has proved to be quite satisfactory
for many designs. However, later design requirements, such as those
for certain low pass and high pass wave-filters in carrier systems, have
demanded, principally from an impedance irregularity standpoint, that
the resistance terminal characteristic be more nearly constant and
extend still farther toward the critical frequencies than is possible with
M-type terminations so as to include in this manner a larger part of the
2 "Transmission Characteristics of Electric Wave-Filters," O. J. Zobel, B. S. T. J.,
October, 1924.
^ See page 17 of paper in footnote 1.
286 BELL SYSTEM TECHNICAL JOURNAL
transmitting bands. A study of this general problem has recently been
made, the results of which were presented in two papers both of
which appeared in the same issue of this Journal.^ The terminal
transducers there described consist of simple non-uniform ladder type
structures whose series and shunt impedances are each arbitrarily
proportional to the corresponding impedances of the "constant ^"
wave-filter and of two-terminal reactance networks added in series
or in shunt at the terminating end to complete them. A transducer
of this kind practically satisfies the ideal conditions in the transmitting
bands, but it does not have a standard image impedance in the atten-
uating bands as is desired here. Because of the latter fact, transmis-
sion loss calculations can not be made as readily as in a composite
wave-filter.
This paper gives the solution of the terminal wave-filter impedance
problem by the logical extension of the use of the general systematic
methods of derivation which had led to the derivation of il/-type
sections, and the use of composite wave-filter principles. The solution
is obtained in two naturally related steps which are, first, the derivation
of sections having mid-point image impedances which are desirable as
terminal wave-filter impedances and, second, the formation of terminal
wave-filter transducers having these image impedances at terminals.
A brief outline of these steps will be given here before proceeding with
the details.
The first step, the derivation of suitable terminal sections, is based
upon the use of two fundamental operations for deriving structures
already mentioned which are applicable to any ladder type network.
One of these, the mid-series derivation whose operation will be desig-
nated symbolically as Di{s), derives from any prototype a more general
ladder type structure whose series and shunt impedances are such
functions of the prototype impedances and of an arbitrary parameter, s,
that its mid-series image impedance is identical with that of the
prototype and thus independent of 5. Its mid-shunt image impedance
is, however, a function of this arbitrary parameter, where < 5 ^ 1,
and is thus more general than that of the prototype at the correspond-
ing termination. The other operation, the mid-shufit derivation desig-
nated as Di{s), derives from a prototype another more general structure
whose mid-shunt image impedance is identical with that of the proto-
type but whose mid-series image impedance depends upon 5. If both
of these prototypes, not necessarily the same, have identical transfer
constants, then both derived structures having the same value of
^ "A Method of Impedance Correction," H. VV. Bode, B. S. T. /., October, 1930.
"Impedance Correction of Wave-Filters," E. B. Payne, B. S. T. J., October, 1930.
ELECTRIC WAVE-FILTERS 287
5 will also have identical transfer constants which are functions of s.
At the limiting value of the parameter, 5=1, each derived structure
becomes identical with its prototype. The reason for the use of 5 as
the general parameter instead of m, as in previous papers, is to permit
it to take on without confusion a succession of values including m, as
will be seen.
Beginning with the "constant fe" wave-filter of any class as the
initial prototype, these two operations are performed alternately on
successive structures, which results in producing two different sequences
of wave-filter structures, depending upon which of the operations is
first used. These wave-filters are all of the same class and contain
successively more and more elements. In Sequence 1 (see Fig. 4) the
first operation is Pi(w), then D-iim'), Di{m"), etc., the parameters being
taken in succession as 5 = w, m' , m" , etc. In Sequence 2 (see Fig. 5)
the first operation is Di{m), then Di{m'), D^im"), etc., with the same
succession of parameters as before. Since at each derivation another
single parameter is introduced, each successive structure of either
sequence has one more arbitrary parameter than the preceding struc-
ture and the number of arbitrary parameters in any structure is equal
to the number of alternate operations performed to obtain it from the
' ' constant k ' ' wave-filter. Now every section has one mid-point image
impedance which is a function of all of its arbitrary parameters. Hence,
this whole process is effectively one for obtaining a structure with an
image impedance which contains any desired number of arbitrary
parameters. The first derived structures in both sequences are the
pair called M-types having the parameter m. The second derived
ones will be called the pair of il/M'-types with parameters m and m' ;
the third, the pair of J/.l/M/"-types with 7n, m' and m" \ etc. Each
successive pair can have a more nearly constant resistance impedance in
all transmitting bands than the preceding pair because of one additional
parameter in the image impedance functions. The two members of a
pair have identical transfer constants and either member can be
obtained from the other, as inverse networks of impedance product B?.
While the derived structures are wave-filters having the same
transmitting bands as the "constant ^" wave-filter, their propagation
characteristics are otherwise more general. However, no different
propagation characteristics are obtained in the successively derived
structures than are possible with the first derived or M-types since
all these derived structures have potentially identical transfer con-
stants, the transfer constant of any structure being dependent upon
its parameters only in their product. A simple relation is given
here between these parameters, the frequencies of infinite attenuation
288 BELL SYSTEM TECLINICAL JOURNAL
and the critical frequencies belonging to any of these derived sections;
there is a slightly different relation for each of the four general groups
into which all the different classes of multiple band pass wave-filters
may be divided. The MM'-types, etc., are structurally more com-
plicated than ilf-types and therefore have preferential value from an
impedance standpoint primarily.
The second step of this solution, the formation of terminal wave-
filter transducers, is related to the first step. The method of deriving
sections which possess desirable terminal image impedances furnishes
through the successive operations the necessary means whereby the
final impedance section can be joined to the standard "constant ^"
wave-filter without impedance irregularity. There are two such
general transducers, the series terminal transducer which connects to
the standard mid-series image impedance and the shunt terminal
transducer which connects to the standard mid-shunt image impedance.
Obviously the series terminal transducer is obtained from the wa\-e-
filters of Sequence 1 and is formed by connecting in tandem mid-
half sections of successive derived structures, beginning with the
series .l/-type and ending in the one having the desired image imped-
ance. At each junction point, always between dissimilar sections, the
image impedances are identical and in every case it is possible to merge
the adjacent series or shunt impedances, thereby considerably reducing
the total number of elements in the entire network. This composite
wave-filter has the same number of dissimilar mid-half sections as there
are arbitrary parameters in the final image impedance function and the
sections are functions of one or more of these same parameters, con-
taining in succession m, m and ni' , m and m' and m", etc., the final
terminal section containing all parameters. The image impedance at
one end of this transducer is entirely independent of all these parame-
ters, being equal at any frequency to the mid-series image impedance
of the standard "constant ^" wave-filter; that at the other end depends
upon them all. Fixing the final impedance characteristic determines
all these arbitrary parameters and therefore all the sections making up
the transducer. The propagation characteristics of these sections,
while similar in form, are all different in frequency placement, being like
those of il/-types having successive parameters equal to the products m,
mm', mm'm", etc. Since m, m', m", etc., are each less than unity, these
products form a decreasing sequence. As a result, the attenuation
peaks of successive sections are progressively nearer the critical fre-
quencies and their combination builds up desirable attenuation
characteristics.
The shunt terminal transducer is obtained in an exactly similar
ELECTRIC WAVE-FILTERS 289
manner, but uses the wave-filters of Sequence 2 and begins with the
shunt .l/-type.
Any pair of these transducers ha\ing the same number and values of
the parameters have identical transfer constants; moreover, either
netw^ork might be obtained from the other, as inverse networks of
impedance product R'-.
Theoretically, with dissipation neglected, the solution of the terminal
wave-filter impedance problem, as outlined above, can be carried to any
degree of approximation desired toward a constant resistance terminal
image impedance in all transmitting bands. Practically, however, it
is here found unnecessary to go beyond the MM'-types which follow in
sequence directly after the well-known J/-types and are thus com-
paratively simple extensions. They meet the desired impedance ideal
well and are in this respect a considerable improvement over the
J/-types just as the latter are an improvement over the "constant k"
wave-filter, as we might expect. By a proper choice of the parameters
m and m' it will be shown later that the J/J/'-types can be made to
have image impedances which are equal to the same constant resistance
within 2 per cent over the greater part of all transmitting bands. In
low pass and band pass wa\'e-filters this nearly constant resistance
extends over a frequency range which is approximately equal to 96
per cent of the theoretical band width. Similar characteristics apply
to wave-filters of other classes. Such a range includes all of a trans-
mitting band except a small region next to each critical frequency
where, how^ever, the wave-filter attenuation makes it practically
useless for transmitting purposes. Each terminal transducer would
then be a composite wave-filter made up of a mid-half section of the
associated M-type of parameter m and a mid-half section of such an
MM'-type of parameters m and m'. While, as already stated, the
Jl/-types and MM'-types have potentially the same propagation
characteristics, the particular values of the parameters m and m'
chosen from the impedance standpoint give attenuation peaks which
in these J/-types are farther away from the critical frequencies, and in
these MM'-types nearer, than in an J/-type of parameter m = .6,
which is generally desirable. Two such fixed designs '" are given here
for connection to the "constant ^" wave-filter of any class at mid-
series or at mid-shunt, respectively. The particular forms these take
^ The reader should keep in mind that such a terminal wave-filter network is
itself a true composite wave-filter of the same class as the standard or "constant k"
wave-filter. Its image impedance at one end is the same as a mid-point image
impedance of the standard, while that at the other end is the mid-point image
impedance of the MM'-type which is desired at the terminal.
290 BELL SYSTEM TECHNICAL JOURNAL
in the four most important specific classes, namely, low pass, high pass,
low-and-high pass and band pass, are also shown.
Finally, two by-products obtained from a further use of these fixed
network designs will be added. One is the ready design of networks to
simulate the mid-point image impedances of "constant ^" wave-
filters. The other leads to the design of networks which simulate the
impedances of a loaded line, approximately a low pass wave-filter, over
the greater part of its transmitting band.
It need hardly be mentioned that these terminal transducers may
be used to terminate a lattice or other type of wave-filter which has a
standard image impedance or, vice versa, that of a derived wave-filter
such as the Mill'-type. In this manner the terminal image impedance
can be altered efficiently from one characteristic to another. The
lattice type (zi, S2) is itself a symmetrical structure.
The procedure for the design of a wave-filter network to meet
specific requirements may even begin with the choice of terminal wave-
filter impedance characteristics, which are physical and not in general
the same at both ends. The terminal, or reflection, losses due to
resistance or other known terminating impedances would thus be
definitely known. With these taken into account the internal part
would be designed using any type or types so as to fit in between the
chosen image impedances without impedance irregularity, as in a
composite structure, and give the remainder of the desired transmission
characteristic.
Part 1. Derivation of Wave-Filters Which Possess Desirable
Image Impedances
1.1 General Ladder Type Structure
Of the three simple general types of recurrent or iterative structures,
the ladder, lattice and bridged- T types, only the ladder type which has
alternate series and shunt impedances, Si and S2, respectively, has two
different image impedances per periodic interval and these are Wi and
W'2. at the two mid-points, mid-series and mid-shunt. The ladder type
can therefore be separated on the image basis into either of two kinds
of symmetrical sections with two pairs of terminals, mid-series or mid-
shunt sections, or into one kind of dissymmetrical section, a mid-half
section. The existence of two different image impedances for a section,
the general property of all mid-half sections, is a necessary condition
for the proper combination of mid-half sections of different related
structures to give the desirable terminal impedance results obtained in
this paper. Definitions of these three kinds of sections which have
been considered in previous papers will be reviewed here.
ELECTRIC 1 1 VI T 'E-FIL TERS
291
A mid-series section is that part between the mid-point of one series
impedance Zi and the mid-point of the next series impedance. It has
the three impedance branches hzi, Zo, and i-i and has the structure of a
T-network. Its image impedance at each end is the mid-series image
impedance Wi.
A mid-shunt section is that part between the mid-point of one shunt
admittance I/S2 and the mid-point of the next shunt admittance. It
-o-/\AAr-o-
Midy-Ahw)iL xjCeniiKLtwrYv
Fig. 1 — -Fundamental derivations.
< 5< 1.
has the three impedance branches 2^2, Si. and Izi and has the structure
of a TT-network. Its image impedance at each end is the mid-shunt
image impedance W^. Both of the above symmetrical sections have
the same transfer constants, 7", as we should expect since both sections
represent one full interval of the ladder type structure.
2'>2 BELL SYSTEM TECHNICAL JOURNAL
A mid-half section is that dissymmetrical part between the mid-
point of one series impedance and the mid-point of the next shunt
admittance, or vice versa. The image impedances at the two ends
are, respectively, W\ and Wi, or vice versa. Its transfer constant is
one-half that of a full section, mid-series or mid-shunt. Obviously,
two mid-half sections when connected with like image impedances,
Wz or Wi, adjacent, will form a mid-series or mid-shunt section,
respectively.
Well-known formulas for the transfer constant, T, of a full section
and for the mid-series and mid-shunt image impedances, Wi and W^,
are
cosh T - cosh (^ + i5) = 1 -f 1^ = 1 + 2{U+ iV),
Wi = VziZ2 + W = VsiZaVl + U+iV,
and
Z1Z2 VZiZ2 21Z2
VziZ2 + W Vl+ U+iV w, ^ ^
where
4Z2
Such a general structure is illustrated in the upper part of Fig. 1.
1.2 Fundamental Derivations
1.21 Mid-Series Derivation by Operation Di{s)
From any ladder type network Zi, z-2 it is possible to derive a more
general one Zi{s), z^'is) which has the same mid-series image impedance
Wi as the prototype, but a transfer constant T{s) and a mid-shunt
image impedance W-iis) which are functions of an arbitrary parameter
5. This operation, denoted as Di{s), is specified by the mathematical
and physical relations between the series and shunt impedances of
the derived network and those of the prototype, namely,^
Zx\s) = SZu
and (2)
where < .y ^ 1 for a physical structure. At the limit s -= \, it
reduces to the prototype. (The superscript "prime" refers to the
case of mid-series equivalence.)
6 See footnote 3. Also U. S. Patent No. 1,538,964 to O. J. Zobel, dated May 26,
1925.
ELECTRIC WAVE.^FILTERS 2<>3
These relations give for the derived structure in terms of its proto-
type and parameter s
cosh T{s) = 1 + 2{U{s) + iV{s)),
Wi = Wu
and
W,{s) = W,[_\ + (1 - .-)(t/+ /F)], (3)
where
s'-iU+iV)
U{s) + iV{s) =
1 + (1 - s'KU+iV)
By the above operation a new image impedance W'y{s) has been
obtained which is more general than the mid-shunt image impedance
of the prototype.
1.22 Mid-Shunt Derivation by Operatioti D-2(s)
From any ladder type network Zi, S2 it is possible to derive a more
general one Zi"{s), z^"{s) which has the same mid-shunt image imped-
ance W2 as the prototype, but a transfer constant T{s) and a mid-
series image impedance W\{s) which are functions of an arbitrary
parameter s. This operation, denoted as 1^2(5), is specified by these
mathematical and physical relations between the derived network and
its prototype
1
^1 \^}
SZl As
1-s^''
z-;'(s)
1
Z-2,
s
and (4)
where < 5 ^ 1 for a physical structure. At the limit 5 = 1, it
reduces to the prototype. (The superscript "second" refers to the
case of mid-shunt equivalence.)
From these relations it follows that the derived structure has
cosh T{s) = \ + 2{U{s) + iV{s)),
Wi
Wi{s) =
1+ (1 -s'){U+iV)'
and (5)
W2 = W2,
where
TU^M -irf^ S%U+iV)
U{s) + iV{s) =
1+ (1 -s'){U+iV)
294 BELL SYSTEM TECHNICAL JOURNAL
This operation gives a new image impedance Wi{s) which is more
general than the corresponding one of the prototype.
The derived structures represented by formulas (2) and (4) as well
as their common prototype are given in Fig. 1. A comparison of
formulas (1) to (5) shows that for the same value of the parameter 5
both derived networks have the same transfer constant T{s) and that
z,'{s)z^"{s) = z,"{s)z.2'{s) = WiWo = Wi{s)Wo(s) = z^z..
Thus the series and shunt impedances of one derived structure are
inverse networks of impedance product ZiZo of the shunt and series
impedances, respectively, of the other one derived from the same
prototype, Si, Zo- vSimilarly, the pair of image impedances Wi and W2
and the pair Wi{s) and W^is) are inverse impedances of this same
product. In fact, either infinite structure might have been obtained
from the other as such an inverse network; the transfer constants of the
two would then necessarily be identical for the ratio of series to shunt
impedance would be the same in both.
1.3 "Constant k'' Wave-Filter, The Initial Prototype
The "constant k" wave-filter of any class, that is, having any
preassigned transmitting and attenuating bands, is a reactance
network of ladder type whose product of series and shunt impedances,
and therefore iterative impedance k of the corresponding uniform line,
is a constant independent of frequency. Putting k equal to the
resistance R of the line or impedance with which the wave-filter is
normally to be associated, we have
ZikZ2k = k^ = R^ = a. constant.
Here and in what follows the additional subscript k implies a relation
to the "constant ^" wave-filter.
When there is dissipation in the reactance elements, the above
relation is strictly satisfied by requiring that the coil dissipation
constant, (/, and the condenser dissipation constant, d', be equal for
each pair of inverse network elements. For example, when d = d'
{d + i)2irfLik _ Lik ^ D2
{d' + i)27rfC2k C.U
There are several reasons for choosing the "constant ^" wave-filter
as the initial prototype.
1. Its structure and method of design for any class is definitely known. ^
7 See footnote 1. Also U. S. Patent No. 1,509,184 to O. J. Zobel, dated September
23, 1924.
ELECTRIC WA VE-FILTERS
295
2. It has both standard image impedances, each of which passes
through the same cycle of values in all transmitting bands.
3. Each il/-type or wave-filter of higher order derived from it can
have an improved impedance characteristic which is the same
in all transmitting bands.
4. The assumption that its impedances Zik and z^k are general in the
analysis makes the results independent of any particular class
of wave-filter and hence applicable to all classes.
5. This method of analysis sorts out certain valuable properties which
are common to all classes by treating known groups of meshes,
Zifc and S2fc, as units, thereby eliminating the necessity of
considering each individual mesh which may be present in the
interior of Zik and z^u of any particular class.
It will be appreciated by the reader that the difficulties of the problem
for one of the higher classes of wave-filters are thus greatly reduced
over what they would be if each mesh had to be taken into account, as
might be required by other methods.
-o-Y^\/-o-
2 ^/A/ 2 '^ikj
-o-Y/V-o O o-IS/\f\rO-
:^2/v
Kk. 2^2^? W.vt^'^zh^
Fig. 2 — -"Constant k" wave-filter, the initial prototype;
ZikZ-ik = kr = R^ = Si constant, independent of frequency.
The "constant ^" wave-filter of any class, shown in Fig. 2, will
be assumed known and is the starting point for obtaining the other
structures which are to follow. It has the formulas
and
cosh r, = cosh {Ak + IB,) - 1 + 2{Uk + iV,),
W,k = i?VH- Uk + iVk = Rik + iXik,
W.k =
R
i?2
diere
Vl+ Uk + iVk
Wi
= R2k + iX-2k;
Tk = transfer constant of a full section,
\Tk = transfer constant of a mid-half section,
Wik = image impedance at a series mid-point,
Wik = image impedance at a shunt mid-point,
Zik ^ / ZlfeV
(6)
Uk+iVk =
4z.,k
2RJ
296 BKLL SYSTEM Tl'.CIIMCAL JOURNAL
and
jR- = Zu-^-ik = ^' = a constant.
It will be noted from these formulas that the transfer constant and
both image impedances of any "constant ^" wave-filter are functions
of frequency only through the variables Ut + iVk, or the equivalent
(su-/27?)- which is a function of su. (It would also hav^e been possible
to use So/.: instead of Zu--) When no dissipation in the elements is
assumed, Su = ru- + ^^u becomes Zi/, = ixv;, a pure reactance, since
then Vik — 0; also Fa = 0. Under these ideal conditions we know that
X\k always has a positive slope with frequency,^ and when the Xik of a
multiple band wave-filter is plotted against frequency it is made up of
negative branches from .vu = — oc to and positive branches from X\k
= to + 3c which lie alternately in succession along the frequency
scale. These branches are defined to correspond with the sign of .tu.
The value of Lh is always negative and ranges continuously with
frequency between the values Uk = and — cc , once for each branch
of Xik. We know also that in a negative branch there is a transmitting
band at frequencies corresponding to values from .vn = — 2R to 0, and
thus from Ui, = — 1 to 0. In a positive branch there is a transmitting
band from xu = to + 2R, thus from Uh = to — 1. A low pass
band is associated with a positive branch which begins at zero fre-
quency while a high pass band is associated with a negative branch
ending at infinite frequency. An internal transmitting band, on the
other hand, has this association with a pair of branches, a negative
followed on the frequency scale by a positive branch, and in reality
consists of two bands which are confluent at .Vu- = 0, i.e., Uk — 0,
where the two branches join. Such a confluent band is formed by the
junction of two bands which occur separately in a wave-filter of higher
class than this "constant k" wave-filter but with the same configura-
tion of elements.
Since all negative branches are similar, as well as all positive
branches, an approximate representation of the frequency charac-
teristics of any "constant ^" wave-filter can be constructed from the
characteristics which belong to each of these two kinds of branches.
It is necessary to consider both a negative branch and a positive
branch since the characteristics of one branch dift'er in their variations
with frequency from those of the other. Dift'erences would naturalh'
be expected from the fact that in formulas (6) which hold for both
branches the variable Uk varies with increasing frequency from
Uk = — =« to in a negati\'e branch and from f/i = to — ^ in a
* See page 5 of paper in footnote 1.
ELECTRIC WAVE-FILTERS 297
positive branch. When Vk = 0, as when no dissipation is assumed,
the formulas become functions of Uk only but contain a certain in-
determinateness regarding the signs attributable to the phase constants
and image impedance reactances of the two branches. This difficulty
vanishes when dissipation is present to give W a value different from
zero, as in a physical wave-filter.
With dissipation such as to preserve the "constant k'' relation it is
readily shown that Vu is negative in a negative branch and positive in
a positive one; that is, Vk has the sign of xu- This follows directly
from the formula
since ru- must be a positive resistance in a passive network. On the
basis of this result it follows from formulas (6) that ^ in a negative
branch
Xik, Vk, Bk and Xik are negative;
X2k and X^k are positive.
In a positive branch these signs are reversed.
The characteristics of two such representative branches are shown
in Fig. 3, joined as they would be to form an internal transmitting
band. The scale of abscissas is Uk rather than frequency in order to
be general, and Uk varies in going from left to right from — oo to for
the negative branch and from to — qo for the positive branch. In
this way a movement along the abscissa-axis from left to right always
corresponds to an increase in frequency. A translation from the Uk
to the frequency-scale can be obtained in any particular case through
the known relationship between Uk and frequency. Such a translation
would be equivalent to a variable expansion or contraction of the above
characteristics parallel to the abscissa-axis. The effects of dissipation
on the different characteristics are indicated by broken lines and show
a rounding-off of abrupt changes. Here, for convenience, it was
assumed that Vk = .01 Z7a- in a negative branch and Vk = — .Olt/i in
a positive branch. If each pair of characteristics Is considered as
separated by an imaginary line perpendicular to the Z/t-axis at
Uk = 0, then a comparison will yield the statement that corresponding
pairs of Ak, Rik and R^k are images of each other with respect to such
lines, while pairs of Bk, X^k and X^k are images but also opposite in
sign.
^ See also page 577 of paper in footnote 2.
298
BELL SYSTEM TECHNICAL JOURNAL
BcndHire
Xifv
ybvcunchi
ZR
^,-^
m_
L--.
__JU
-;
7
-1
^.
ixmmjchi .
^ZR-
'^ih
^lK=^l/V+^'^/ll
^
-/ ^^"0
a.
Fig. 3 — Characteristics of "constant k" wave-filters.
(Broken lines indicate the efifects of dissipation.)
ELECTRIC WA VE-FILTERS
299
1.4 Sequence 1
As already stated in the Introduction to this paper, the successive
wave-filter structures of any class which comprise Sequence 1 are
derived from the known "constant ^" wave-filter taken as the initial
prototype by performing in succession the operations Di(m), then
D^im'), Di(m"), etc. They may be considered as wave-filters of
higher and higher order since they contain a greater and greater
number of arbitrary parameters. The parameters of the alternate
operations Di(s) and D2{s) are in the order oi s = m, m' , m" , etc.
The small letter m with superscripts is used as the notation for all
the parameters in order to denote their association with "mid" of mid-
point impedances, since mid-points are under consideration here in
ladder type networks. Where the initial prototype is the "constant k "
wave-filter, as it is here, I have used a terminology for the derived
structures whose basis is the capital letter M with superscripts to
correspond with those of the associated small letter parameters.
Thus, I have shortened the expression "mid-series derived, parameter
m ladder type" to "series .l/-type"; similarly for the other structures.
"jCormlafrd, h"^ SatJbc^ M-yfypB SMimh M M-^-pe Seri£^M MM-ytype
_D/m^l
•Sequence 1.
The wave-filters of Sequence 1, so designated, can be expressed
concisely in the following symbolic manner where any part within
brackets represents a ladder type structure. Each operation is to be
performed upon the structure within brackets to its right; therefore, to
obtain the actual series and shunt impedances which result in any
particular case when two or more operations are involved, these
operations would begin at the right with Di{m) on [_k], the "constant ^ "
wave-filter.
"Constant ^" = [k~\,
Series i/-type = Di{m)[k'],
Shunt MM'-ty^e = D2{m%Di{m)\ik']'],
Series i/MM/"-type = D,{m")lD.{m')lD,{m)[_k']']'], etc.
(7)
A diagram which illustrates this process and gives as well the
notation of the resulting image impedances in the successive structures
300 HELL SYSTEM TECHNICAL JOURNAL
of Sequence 1 is shown in Fig. 4. Each rectangle represents a wave-
filter of ladder type having the two mid-point image impedances
as indicated. The operation symbol between each succeeding pair of
rectangles shows what operation has been performed and the arrow
points towards the derived structure of higher order, being placed in
line with the image impedances which are identical for the pair. Thus
it is seen that each derived structure has one identical and one more
general image impedance than the preceding structure. In the
sequence the new image impedances appear alternately at mid-series
and mid-shunt points, beginning with the latter here.
The series and shunt impedances of the different structures which
become more and more complicated with increase in parameters are
derived by performing the above operations but their detailed con-
sideration will be deferred to a later point.
The transfer constants of the various members of this sequence are
found by carrying out the proper operations based upon formulas (3),
(5) and (6) and can be expressed by one formula, namely
cosh 7,(g) = 1 + YTir^^TKfATTi^ ' ^ ^
where g = 1, w, mm', mm'm", etc., in a decreasing sequence.^" The
value of g for the structure of any order is equal to the product of all
of its parameters, the first value above, g = 1, being that of the
"constant k'' wave-filter. This is, for example, because by (3)
^ 1 + ( 1 - m-m m" ) ( Uk+i Vk)
The image impedances in Sequence 1 which are derived in a corre-
sponding manner have these formulas.
W.2k{m) = T/ro,[l+a(f/;t+iF,)],
^'^^â– ("'' "' ) ^ \:i+a'im+m)T ' ^^^'^
, ,. W,ll+a(U,+iV,)T^+a"(U,+ iV,):\
IT .A.(m, ni , m ) ^ [l+a'(t/.+ nO,.)]
i» Computations for the transfer constant can be made accurately from formulas
for cosh-i (x + iv) given in Appendix III of the paper " Distortion Correction in
Electrical Circuits with Constant Resistance Recurrent Networks, O. J. Zobel,
B. S. T. /., July, 1928.
ELECTRIC ir.l VE-FILTERS
301
where
4 *>
(7 = 1— ni-,
a = I — nrni
111
a" = \ - nrm'-m"\ etc.,
in an increasing sequence approaching unity. Wxk and Wu- are the
"constant ^" image impedances of formulas (6). The continuation of
this series of image impedances is quite obvious, a new factor appearing
alternately in the numerator and in the denominator.
Each factor in the numerator gives the image impedance a resonant
point in an attenuating band where the image impedance is a reactance
and Uk < -1; that is, at Uk = —l/a, or -1/a", etc., neglecting
dissipation with Vk = 0. A factor in the denominator gives an anti-
resonant point; at Ui: = —1/a', etc. Since a' lies between a and a",
etc., these resonant and anti-resonant points alternate as in a general
reactance network. Only the resonant or anti-resonant point due to
the new factor added coincides with the point of infinite attenuation
in the corresponding new structure, as may be seen upon comparing
formulas (8) and (10), neglecting dissipation. These properties out-
side a transmitting band may or may not be desirable in certain kinds
of circuits. They are of importance when considering terminal losses
in an attenuating band, as in Section 2.6.
1.5 Sequence 2
Here the derived structures are obtained by performing in succession
the operations D^im), then Di{m'), Di{m"), etc., where the initial
"Ccrriblaml hT Shujtil M-Jyyfxe S^aa^ MMzty^ Shwrib MMM-iype
]]^(7n)
Dj(7ri)
R(-^^)
Fig. 5 — Sequence 2.
prototype is the "constant k" wave-filter. Using the same notation
and terminology as before, the wave-filters of Sequence 2 when ex-
pressed symbolically are
"Constant ^" = |^^^,
Shunt if- type = D2{m)[k'],
Series J/J/'-type = D,{m')[_D.{fu)[kJ],
Shunt .l/,l/M/"-type = D;{m")lD,{m')[_D-,{m)[_k'J]'], etc.
(11)
302 BELL SYSTEM TECHNICAL JOURNAL
A corresponding diagram which illustrates this process is that of
Fig. 5.
The transfer constants of these wave-filters are also given by formula
(8) which includes (9).
The image impedances in Sequence 2 are
. W,,{_\+a'{U,-\-iVu)'] (12)
where a, a', a", etc., have the same values as in (10).
1.6 Relations Between Sequence 1 and Sequence 2
Carrying through operations for the determination of the structures
of the series and shunt impedances in these wave-filters, the following
results are found:
a. Each pair of structures of the same order in the two sequences is a
pair of inverse networks of impedance product F?.
That is, if the series Tlf-type has the series and shunt impedances
Sifc'(m) and z^kim), and the shunt il/-type Zv/'{m) and So/'(;;0, the
inverse network relations are
z,,'(m)z2,"(m) = z,u"{m)z,u'{m) = R\
For the il/il/'-types, using similar notation,
su'(w, m')z2k"{ni, m') = zn"{m, m')z2k'{m, m') = R',
and so on for the higher order pairs. Consequently, one structure of
each pair might be obtained from the other as such an inverse network.^^
b. The transfer constants of both structures of a pair are the same.
This result would come from the inverse network relations which give
both structures the same ratio of series to shunt impedances, a ratio
which determines the transfer constant. It has already been found in
formula (8) where the value of g is the same for both structures of any
order.
11 The structures indicated or to be shown in detail in Sequence 1 and Sequence 2
can be generahzed as ladder type derivations from any initial prototype Zu s^. This
is done by a simple replacement of su- and Cji by Zi and So, respectively; of K- by the
product 3122; and by the omission of the subscripts, k, throughout.
ELECTRIC WAVE-FILTERS 303
c. The series and shunt image impedances of a pair are inverse nehvorks
of impedance product E?.
Such results would also follow from (a) above together with the
consideration of mid-point terminations. They are verified by com-
parison of formulas (10) and (12) which give
Wu:W-ik = Wi,(m)W2,(?n) = Wi>c(m, m')W2k(m, m')
= Wik(m, m', m")Wik{m, m' , m") = • • • = R-.
d. Both image impedances of either MM'-type, or of either one of a
higher order pair, may be adjusted dependently without changing
its transfer constant; the ratio of the two image impedances is
fixed when the transfer constant is fixed.
This can be seen from the fact that the transfer constant depends upon
the parameters only in their product, g, and from the formulas for two
consecutive impedances in (10) or (12).
1.7 M-Type Wave- Filters
These are the wave-filters of the first order in each sequence and
contain one arbitrary parameter, m. Although they are quite well-
known, it is necessary to include them here for the sake of continuity
and because of the fact that they are to be used later.
The series .l/-type has the formulas
Zik(m) = mzik,
,, . 1 — m- 1
cosh r.(;;0 = 1 + ^^'iUk + iV,]
(13)
1-f (1 -m''){Uu + iVk)
and
Wik = i?Vl+ Uk + iVk,
^ i^[l+(l-^n-)(^. + .-F.)]
Vl+ Uk + iVk
In the shunt .l/-type
zik"{m) = — -^ ,
mzik 4 m
1 — m''
.V(m)=^s,„ (14)
cosh Tk{m) = same as in (13),
i?Vl+ Uk-hiVk
W,k{m) =
[1+ (1 - w2)(f/, + iF,)J'
304
and
BKLL SYSTEM TI.CIIMCAL JOURNAL
W;u =
R
VI + U,+ iV,
In the above < m ^ \. At the limit m = \, the two structures
reduce to the "constant k" wave-filter; also IFi/,(w = 1) = Wu- and
W2k(m - 1) = IF.,.
A mid-half section of each of these wave-filters is shown in Fig. 6.
It is to be remembered that the transfer constant of a mid-half section
is one-half that of the full section given in the formulas.
MlcC-JwU/ Aeam- M-Jyype
z
-o-VvAv-o-
m
Jh
27n ^/h
•A 7
•772/ *^Zh
K^cm)
'2h
MlcC-Jmlf Ahunb M-Jyy^e
TTh-y
2 ^iK
zm
i-7n^ ^zhj
K,
^zh
Zo
7rh*^zh
w,m
Fig. 6 — -Alid-half sections of M-type wave-filters.
To illustrate the propagation and impedance characteristics of
ilf- types, as in Fig. 7, the parameter was taken to have the value
m = .6. The attenuation constant has one maximum just beyond
each critical frequency, where Uh = — 1/(1 — nr) = — 1.5625, and
in this particular case the image impedances shown have the fairly
constant resistance values over a large part of each transmitting band
to which reference has been made. With other values of m there may
ELECTRIC WAVE-FILTERS
305
or may not be in the range from Uk = to — 1 one maximum for
W],k{m) and one minimum for W^him). The image impedances at the
other mid-points are independent of m and are identical with those of
the "constant ^" wave-filter already shown in Fig. 3.
Af^m
-1
Ifr)
-I
(frJ
u,.
Tt â–
^
Bi^lm]
~\ ~' ^
-i
u^
^^ n
Fig. 7 — Characteristics of M-type wave-filters;
m = .6.
{Wxk and Wik are illustrated in Fig. 3. Broken lines indicate the effects of dissi-
pation.)
1.8 MM'-Type Wave-Filters
As wave-filters of the second order in each sequence they have two
parameters, m and m' . Their series and shunt impedances are derived
by means of the single operations with parameter m' performed in the
regular manner upon the iU-type structures as prototypes which have
the formulas (13) and (14).
306 BELL SYSTEM TECHNICAL JOURNAL
Formukis for the series il/i/'-type are
Si//(w, ni') = z ,
+
mm'zik 4inm'
r> Zok
nr
cosh Tj,{))i, m') = 1 +
4m' '^'' w'(l - m-')
2mhn'\Uu + iVk)
1 + (1 - ni^nr){Uk + iVt)
and
R[\ + (1 - mW^)(;7, + zF,)]
W-ik{m, m') =
[1 + (1 - W')(t/, + iF,)]Vl + f/;t + ^'F,
where < w ^ 1, and < An' ^ 1.
As a limiting value, W^kijn, m' = 1) = W^k-
For the shunt MM'-type
Zik"(m, m') =
' +
mm'zik m'{\ — m^) . Am'
m{\ — m' ) m{\ — m' )
Z'lk {m, m ) = —. r -u + ■/ -2/:,
Anim mm
cosh Tk{m, m') = same formula as in (15),
i?[l + (1 - m'){Uk + iVk)yi + Uk + JF,
[1+ (1 - 7nhn'^)(Uk+ iVkU
TFu(w, ^;i') = ,-.,.. ., /2
and
Vl+ f/. + ^'F,
where as before < ;;z ^ 1, and < m' ^1. A limiting value here
is WuOn, m' = 1) = TFu-
The J/.l/'-type wave-filters have structural designs which can be
inferred from their respective mid-half sections of Fig. 8; they may
have characteristics such as illustrated in Fig. 9 where the parame-
ELECTRIC WAVE-FILTERS
307
ters are m = .7230 and ;;/' = .4134; the reason for this particular set
of vakies will be explained later. The transfer constant is the same as
that of an J/-type of parameter equal to the product mm' = .2989.
With other values of /;/ and ;;/' the image impedances Wik{m, m') and
W-ik{m, m'), which in the transmitting bands are pure resistances if
dissipation is neglected, can be given a variety of characteristics as is
apparent from their formulas. In fact their physical possibilities can
WJ'^^ mitinBy
zm(t-rnJV -r ,
Wouf^''^')
zmii-Trv^) ^ih mO'TTV^) ^zh
-o-/V\A^
mrrv-y
z ^?h
M,(m)
hK,
ZTmn^ih
Kjrn,7rv)
TfFi
''^zh
Fig. 8— Mid-half sections of ilfilf'-type wave-filters.
then be described by the following statement. In the range from
[/a,- = to — 1 the characteristic corresponding to the positive ratio
y = Wu{m, m')JR = R/W2k{'m, m') may have no maximum or mini-
mum, one maximum, or one maximum and one minimum; at Uk = 0,
y = I and at Uk = -1,3' = 0. All of these structures which have
the same value of the product g = mm', have the same transfer
constant. Thus, it is possible to keep the transfer constant fixed and
vary the image impedances.
No structures of any higher order will be worked out here in detail
since for all practical purposes the .l/M'-types just considered will be
308
HELL SYSTEM TECHNICAL JOURNAL
found capal)le of meeting the ideal impedance requirements. If
desired, the structures for the MM'M"-types and higher orders can
easily be derived by the regular operations indicated. In them some
slight reductions in the number of elements can be made because there
are then three or more similar impedances in one branch.
A^(mpn')
-I
-I
Uk
y\,j^cm,Trii
i|X,;t,('W,7r2''j
^zh.
\/
Characteristics of il/M'-type wave-filters;
m = .7230, m' = .4134.
Broken lines indicate the effects of
(iru(/») and Wik{m) are illustrated in Fig. 7
dissipation.)
It should be quite obvious that a wave-filter of any order reduces
to the "constant k'" wave-filter when every one of its parameters
reaches its limiting value, unity.
1.9 Frequency Relation in the Attenuation Characteristic of an
M-Type or Higher Order Wave- Filter of Any Class
The attenuation characteristics of J/-type and .l/.l/'-type wave-
filters which have been illustrated in a limited frequency range show
ELECTRIC WAVI'.-EI ITERS 309
that when dissipation is neglected there is infinite attenuation at some
frequency within each branch of Xik. Formula (8), when Vk — 0,
gives in the attenuating bands where C7/. ^ — 1
cosh Au{g) =
1 + ^^'^'■•
1 + (1 -ewu
(17)
in which g = ;;/, uim' , mm' m" , etc., for the .^/-types and higher orders.
The critical frequencies occur where the attenuation constant becomes
zero, i.e., at Vk = — 1, while the frequencies of infinite attenuation
occur where it becomes infinite at Vk = — 1/(1 — g^)- Since, when
Vk^O, (si/,/2i?)- = Vk, we have the following results:
At critical frequencies /o, /i, etc.,
zu- = ± i2R. (18)
At frequencies of Infinite attenuation, /oco, /loo, etc.,
^u-=±-y££=, (19)
^l - r
the number of such frequencies being equal to the number of critical
frequencies.
A very simple relation has been found between these two sets of
frequencies in the case of any multiple band pass :l/-type or higher
order wave-filter. Such a relation is given here for each of the four
general groups into which all classes of band pass wave-filters may be
divided, each group having n internal bands with or without low pass
and high pass bands.
Group 1. — Low-and-» Band Pass.
f Ox fix ■• * f2nx = , . ,/o/l ' ' ■/in- (-^)
VI - r
Group 2. — n Band-and-High Pass.
fl^hx • • ■/(2n+l)oo = Vl - r/1/2 • • • /^n+l. (21)
Group J. — Low-7/ Band-and-High Pass.
/Ooo/loo • • ■/(2n+l)oo = /o/l * ' * fin+l- (22)
Group 4. — n Band Pass.
/loo/.. • • • hnx = /1/2 • • • /.>„. {li)
For this group there is a further relation but it applies to the
310 BELL SYSTEM TECHNICAL JOURNAL
impedance characteristics. It contains those frequencies in the trans-
mitting bands where all image impedances become equal to R and
where the series impedances belonging to the different orders become
resonant. These resonant frequencies fu, f^r, etc., are the same as
those of Zik\ that is, where Zik = 0. The relation is
fuhr "'fnr = V/1/2 ' ' ' /.„• (24)
It may be noticed that relations (20) and (21) for Groups 1 and 2 are
the only ones which depend upon the parameter g. The proofs of all
these relations are to be found in Appendix I together with certain
reactance frequency theorems.
Part 2. Formation of Terminal Wave-Filter Transducers
2.1 General Design Method
In the Introduction of this paper the method of forming the two
general kinds of transducers under consideration has been quite fully
discussed. Hence, only a brief repetition will be made here.
The series terminal transducer is designed for connection to the
standard mid-series image impedance, Wn, and is formed by con-
necting in tandem an arbitrary number of single mid-half sections of
successively derived structures in Sequence 1, beginning with the
series il/-type. The image impedances are identical at each junction
and adjacent series or shunt impedances can be merged. The number
of arbitrary parameters in the final image impedance function is equal
to the number of mid-half sections which have been so united. This
impedance characteristic is then fixed to give a desired physical result,
whence the parameters of all intervening mid-half sections are like-
wise fixed. The attenuation peaks of successive sections are nearer
and nearer the critical frequencies.
The shunt terminal transducer for connection to the standard mid-
shunt image impedance, W^k, is designed in a similar manner from the
wave-filters of Sequence 2, beginning with the shunt J/-type.
From a theoretical standpoint the more mid-half sections used in
this composition to obtain a desired constant terminal impedance, the
better the possible approximation. The same method of solving for
the parameters can be used in all cases. But, in practice, two sections
appear to be sufficient.
2.2 Transducers Having Tivo Parameters
Proceeding on the above basis the two-parameter structures of
Fig. 10 are obtained. Their formation will be obvious from Figs.
ELECTRIC WA VE-FILTERS
311
6 and 8, taking into account the merging of similar impedances at the
junctions.
Oy&nerML /xedjei^ .ter/minrwL 2rxifri^^
m
o—
w,,
ih
M.M
o
W,.(m,m)
Try y
Z ^ih
—o->\fs[\rO-
7n'0-7n^) ^ zm! y
zml 1-771) v^ih/ rrh(i-7n'^) ^zh/
I-7TV
ZTTbO-
mm'
^ih
Wjj^(rn,,7n)
>^ y
7rhit*m/)*^zh
o 1 I o 1 I o
«
zh
o—
W,^(rrv)
]/\/.,(7nM}
-o
^^K
7n(i-^m ')'y
2 ^'h
zmo+7n') 'y
'771 ^2hi
1-771^ ^zh
77UJ-771'^)
zrrv
Fig. 10 — General terminal transducers.
The transfer constants of both structures are identical being given by
T = h[T,{m) + n{m, ;;/)]. (25)
At their initial terminals the image impedances are respectively the
standard ones, IFu- and W2k, which have the relations
W
R
R =»^ = ^l+^' + ^»''^
(26)
312 BELL SYSTEM TECHNICAL JOURNAL
and at their I'mal terminals the image impedance relations are functions
of ;;/ and tu' , namely,
^ Wn{m., m') ^ R
^ ~ R IVuim, m')
^ [1 + aiU, + iV,)']<\ + Uk + iV, .^j.
[l + a'(t/, + jFA.)] ' ^" ^
where a = 1 — nr, and a' = I — m-m'". Since m and m' lie between
zero and unity, it follows that ^ a ^ a' < 1.
When there is no dissipation in the network elements, Vk = and
all these image impedances are pure resistances in all transmitting
hands. Then the image impedance ratio y is there real and it can be
given a variety of characteristics depending upon the choice of parame-
ters a and a'. For the range Uk- = to — 1, 7 as a function of Ui,
may have no maximum or minimum, one maximum, or one maximum
and one minimum; at Uh = 0, y = I and at U,c = — 1, y = 0.
The parameters corresponding to any such physical characteristic
can be determined from the values of y at two non-zero values of Uic,
where now
_ [1 + a^,]Vl+ U^-
^~ [1 + a't/,]
This, when rewTitten, yields the general linear equation in a and a'
— 11a + va' = %v, (28)
where
« = - t/,Vl + Uu,
V = — ylJk,
and
IV = y — Vl + Uk.
For generality, let the data be
J = ji at {Uk)u
and
y = y-i at (Uk)-!.
Substitution of these values in (28) gives two simultaneous linear
equations in a and a' whose solution is
_ ViW-y — V2IV1
U1V2 - llfVl '
and ^^^^
, UiW-2 — n-fci\
a = •
iliV2. — ll-lVl
ELECTRIC WAVE-FILTERS 313
Then from (27)
m = Vl — a,
and (30)
; i - a'
The maximum and minimum values of y (where dyjdUi: = 0) are at the
two values of Uk
TT -- (^(^ - «') ± ^ ^^'' - '^y - ^^^'(^ + 2a - 2a')
\Miere it is desired to have an especially constant value, y = 1, in
the neighborhood of Uk = 0, the parameters might be determined
from an expansion of the expression for y in powers of Uk- Equating
these coefficients of the first and second powers separately to zero
would give two independent equations from which to derive the
parameters.'-
2.3 Fixed Designs
The primary interest here is to obtain designs in which the final
image impedances are approximately constant resistances equal to R
over the entire useful parts of all transmitting bands. Such imped-
ances require a j'-characteristic which is close to unity from Uk = to
the neighborhood of t//o = — 1. With this objective a few preliminary
trials showed that very satisfactory results are obtained with the
assumed data
yi = I at {Uk)i = - .65,
y.= \ at {Uk)2 = - .90.
Then from (29) and (30) of the previous Section
a = .4773, a' = .9107;
and (32)
m = .7230, m' = .4134.
These values fix the general structures of Fig. 10, giving the specific
ones of Fig. 11 which are made up of definite proportions of the
impedances Su and Z2k of the "constant k" wave-filter of that class,
assumed known. The detailed ^-characteristic of Fig. 12 shows
that in this case there is less than a 2 per cent departure of v from the
constant value unity over the continuous range from Uk = to
'2 A problem of terminal impedance is also included in the paper, "Die Sieb-
schaltungen der Fernmeldetechnik," W. Cauer, Zeitscliriftfiir Aiigeiiuindle Matheniatilc
iind Meclianik, October, 1930, p. -125—433.
314
BELL SYSTEM TECHNICAL JOURNAL
lu/xecC Ae'Tve^ ZerymMiaJL Jjxirmx<j(ucer
.3615 Zjh
— oAW^-o
J6a6Zik W9Zzj,
w,
ih
.2355 Z,h
1957 Z^f,
.m^z,f^
fY^^(m,m'J
.5/10 Zjh,
m.
'ihj
2.766 Z2HJ -7250 Z,}y
6.076Zzh
6.691 Zzf,
Fig. 11 — Fixed terminal transducers;
m = .7230, m' = .4134.
r
\
/,
uz
/
\
\
Ile<
Tl
/,
1.
01
00
/
\
^
\
yy
\
/
^
\
^^
-^
99
â– ^
\
.98
fV,H,m.vi) ^ 71 (m^,7Z30, 7n'=.4i3a)
oz
JnnaQ
wnany
01
/
10 ,â– â– -.
8
6
4
Z
2
v.-
6" "-:
'8 -j\
;'
01
-OZ
Fig. 12 — Detailed terminal image impedance characteristics in the transmitting
bands of lived terminal transducers.
(Broken lines are for dissipation with Vk = ± .01 L>).
ELECTRIC WA VE-FILTERS
315
/
\ /^
3
Ki
'\
/ .
\ /
V
.\
S:
-^
2
i
VA
/
A
1
/
1
/
I
J
u,.
-z
-3
-Tf
.
.- — ^
3
i
n
1
21 j
\l
\z
2
1
//
\
\
z
5
/
A
',
/
//
'/
r
'\
3
'\
2
"
/
Oi
~
' L
4 -
2
3
/',
//
2
\
l/
/ "'
2'.
I r
-2
^, _^'
-3
--;/
Fig. 13 — Transfer constants (T = ^ + iB)^
(1) of fixed terminal transducers,
(2) of comparison transducers.
(A comparison transducer consists of one mid-half section of the "constant /&" wave-
filter and one of either i/-type, where ni = .6. Broken lines are for dissipation with
Vk = ± .01 Uk).
316
BELL SYSTEM TECHNICAL JOURNAL
Uu = — .92 in every branch, which range includes the useful part of a
branch. In low pass and band pass wave-filters this total range
corresponds to 96 per cent of the theoretical band widths. From (31)
there is a minimum y = .9857 at Uk = - .3696, and a maximum
y = 1.0198 at Ui.- = — .8297. Of course, other values of the parame-
ters in this neighborhood would also be quite satisfactory. They
might even be fixed by choosing the frequencies of infinite attenuation
in the two half sections. But the above were taken in order to fix the
final networks.
-/.
-.4
zk/
lb, y
WinffnM) _ R
2b, y= ^L = ^^-^_^^^
, {171=7250, rri^mn)
iMg. 14 — -Image impedance characteristics in the transmitting bands —
(la, 1/)) of fixed terminal transducers,
(la, Ih) of comparison transducers.
(Broken lines are for dissipation with Vu = ± .01 Uk).
The transfer constants of these fixed terminal transducers of Fig.
11 are represented by the general attenuation and phase characteristics
of Fig. 13. Here also are shown the corresponding characteristics
of two comparison transducers, one of which is made up of a mid-half
section each of the "constant ^" and of the shunt .l/-type wave-
filters and has the image impedances Wxk and PFu(w). The other,
made up similarly, has the image impedances Wik and T4^2a(w). In
ELECTRIC WAVE-FILTERS 317
both comparison transducers m = .6, this value of the parameter
giving results which are representative of the best constant terminal
impedances possible in transducers with terminal M-types. (These
comparison networks are identical with the general ones of Fig. 10
in which m = 1 and m' = .6.) Corresponding image impedance
ratios in a transmitting band are given in Fig. 14 where curves la
and lb are characteristics for the two ends of the new terminal trans-
ducers of Fig. 11, while curves la and 2b are those of the comparison
networks. The superior merits of the new transducers can be seen
from Figs. 13 and 14; for in addition to giving improved and prac-
tically ideal terminal impedances they have attenuation characteristics
just outside the transmitting bands which rise more rapidly than those
of the comparison transducers.
By the use of such and other fixed terminal transducers at one or
both ends of a wave-filter network, the flexibility of the composite
method of designing wave-filters is still retained. The transducer
transfer constants and terminal losses due to reflection at given termi-
nating impedances are known in advance. The interior of the com-
posite wave-filter can then be built up of ladder, lattice or other types
of sections so that the desired total transmission characteristic is
obtained. Constant resistance phase networks can also be added at a
resistance termination to help improve the phase characteristic in the
transmitting bands, if necessary.
2.4 Designs for Lozv Pass, High Pass, Low-and-High Pass and
Band Pass Wave- Filte rs
These fixed transducers of Fig. 1 1 may readily be translated into
the particular designs which they assume for any class of wave-filter
with Zik and Zofc known. For low pass, high pass, low-and-high pass
and band pass wave-filters, the four most important classes, the actual
physical arrangements and formulas for the inductances and capacities
have been worked out. As a convenience in reference these designs
are placed in Appendix II where all necessary formulas are given,
making use of Appendix II of the paper mentioned in footnote 1.
Little further discussion will be given here except to add the relations
between Uk and frequency for these different classes, with dissipation
neglected. By this means the characteristics which have been shown
as functions of Uk may be referred to the frequency scale as the
abscissa-axis, if desired in any particular case.
I. — Low Pass
TI,. = - I
Jo
Uk= -{() , (33)
318 BELL SYSTEM TECHNICAL JOURNAL
and xvc is made up of one positive branch.
II.— High Pass
U, = -ij) ^ (^4)
and Xik consists of one negative branch.
III. — Low-and-High Pass
Uk — 7~1 77 7~T2 ' (^^)
/O/I //la /
where j\a = y/o/i, the anti-resonant frequency where Uu = <» and
xifc = CO , For this class xu: has a positive branch from to fxa and a
negative branch from fu to oo .
IV. — Band Pass
C/, = M_ l^-^ - -^ V, (36)
(/2-/l)n f UrI '
where /ir = V/i/o, the mid-frequency or resonant frequency where
Uk = and xu- = 0. Here xik is made up of a negative branch in the
frequency range from to/i,- and a positive branch from/ir to oo .
2.5 Equivalent Structures
Many structures can be obtained which are externally equivalent to
each of the above transducers ; in fact, an infinite number is possible.
That this is so can be seen from a consideration of the general trans-
ducers of Fig. 11, for example. It will not even be necessary to
include the entire networks in this discussion but only the branches
containing three impedances of two kinds, Zik and Zofc- The branch
containing one of Zn- in parallel with the series combination of one of 0u-
and one of Zok may be transformed completely by a well-known formula
into one of Zn in series with a parallel combination of one of z^ and one
of Z'2K- No change in the number of impedance elements results and the
magnitudes are fixed. If, however, an arbitrary part of the original
parallel Zik branch is kept out of the above transformation the final
equivalent structure would have one more Zik impedance and one more
mesh than the original. The proportions of each impedance may
obviously be varied continuously as the arbitrary division is so varied,
thereby giving an infinite variety of magnitudes. This four impedance
structure, equivalent to the original one, reduces at the limits to the
two structures each having three fixed impedances, as we know. A
similar process can be carried out with the shunt branch in the shunt
ELECTRIC WAVE-FILTERS 319
transducer which cxjiitains three impedances. In this case the series
z-21: impedance of this branch would l)e arbitrariK' divided and one part
transformed by another well-known transformation w^ith the parallel
branch in series with it. The final result would be a Zoa in series with a
parallel combination of a z-^u and series Su and So/.-; that is, four imped-
ances but no additional mesh. Here again the magnitudes would
have a continuous range but at the limits with three impedances they
are fixed. Other methods of transformations can be used on the
network as a whole and most of the equivalents have more elements.
As a matter of interest a number of equivalents of the networks of
Fig. 11 will be pointed out, all of which have the same minimum
number of impedances. Starting with the transformations mentioned
above, the latter series transducer has a star of su impedances which
may be transformed into a delta, thereby adding another mesh.
Similarly the latter shunt transducer has a delta of z-^k impedances
which may be given the form of a star which eliminates a mesh. Two
other forms are given as Vi and V^ in Appendix II, being respectively
equivalent to the series and shunt transducers. They are inverse
networks just as are the originals in Fig. 11, In Vi a still further
transformation can be made from a star to a delta of Su- impedances;
in Vo, from a delta to a star of z-zk impedances. The possibility of
obtaining the particular forms Vi and V^ was pointed out by H. W.
Bode. I have derived them directly from the networks of Fig. 11
by a transformation of the major part of each network, using the
simple formulas for the equivalent transducer transformations, re-
spectively 1 and 2, of Appendix III.
The transformation formulas for these latter equivalent transducers
in Appendix III are readily verified by the ordinary transformations
from 2" to TT networks, and vice versa.
In the higher class wave-filters which contain more than one element
in Zik and Soa-, transformations of only parts of Zik and Zik are also possible.
For various other kinds of transformations see footnote 16 to Appendix
III.
2.6 Terminal Losses at MM' -Type Terminations
When the terminal image impedance of a wave-filter is PFu-(/«, m') or
Wok{m, m') and the wave-filter is terminated by a resistance R, there
is a reflection loss at the junction due to the impedance irregularity
which will be called the terminal loss L,n,m'- It is defined by the
relations
T , \R+ Wik{m, m')
2iRWxk{m, m')
R + Wu{m, m')
2^RW2k{m, m')
(37)
320 BELL SYSTEM TECHNICAL JOURNAL
which are exactly analogous to formulas (33) and (34) of the paper
cited here in footnote 2. Thus L,„,m' may be plotted so as to give an
additional chart for use in the method of calculating wave-tilter
transmission losses considered in that paper, which will apply when
there are these kinds of MM'-type terminations. As a convenience a
chart for Lm,m' is given in Appendix IV for the particular values of the
parameters m = .7230 and m' = .4134 already chosen in the fixed
terminal transducers. To take account of dissipation several curves
are shown for each one of which there is a different fixed relation
between Vk and Uk. This chart, being an extension to the former set
of charts, is numbered consecutively with the others as Chart 20. It
shows that the terminal loss at R has two maxima beyond each critical
frequency where Uk = — 1- Their locations correspond to one reso-
nant and one anti-resonant point of WikUn, m') or W^kim, m') in
an attenuating band. Moreover, the position of the first and lowest
maximum coincides with that of the maximum attenuation of the
terminating wave-filter, the MM'-type, while the position of the
second coincides with that of the maximum attenuation of the related
M-type. (An ilf-type termination gives only the first maximum;
an MM'M"-typQ gives three maxima, etc.) The transmission unit,
the Neper, is the same as that which was called the attenuation unit
on the previous charts. The corresponding number of decibels is
obtained by multiplying the number of Nepers by 8.686.
When such a termination is used the interaction loss is practically
negligible.
Part 3. Simulation of Wave-Filter Impedances
So far the two networks of Fig. 1 1 have been considered only from
the standpoint of their use as terminal wave-filter transducers with
desirable propagation and image impedance characteristics. While
this is their major purpose they can have a minor use to be shown
here, namely, as parts of two-terminal networks whose purpose is to
simulate wave-filter impedances where such networks may be desired.
This possibility is suggested by the fact that the image impedances
at the final terminals are approximately equal to a constant resistance
in all transmitting bands which can be simulated at these frequencies
by a simple resistance R. It follows that if each pair of final terminals
is terminated by a resistance R, the impedances at the two remaining
pairs of terminals will be approximately equal to their image imped-
ances, Wik and Wik, respectively, in the transmitting bands. More-
over, on account of the high attenuation of the transducers in the
attenuating bands which reduces transmission through them, the large
impedance irregularities at those frequencies between each network
ELECTRIC WAVE-FILTERS
321
and its terminating resistance R will produce only a small effect upon
the impedances at the other terminals. As a result the latter imped-
ances will be approximately equal to W^k and W-k in the attenuating
bands also. Higher order transducers might also be used.'''
Ajuhjuch /lijirmlated^ W,h,
1,
.36/5 Z,^
— o-VW-° —
1646 Z,h 1-^79 Z^j,
.Z355z,f,l .maz.H,
n
Mlob-hhu/nb ifmp^edojrhce /miAjjovh
.5110 Z;/,
z.
\z.766z2fi,
a.Z8z ZzH,
7Z50Z,j,
6.076Z2h
6.69JZ^^
R
Fig. IS — Impedance networks which simulate the image impedances, W\k and Wik,
of "constant yfe" and related wave-filters of any class.
With this explanation of their origin the general impedance
networks of Fig. 15 have been assembled. One of impedance Zi
simulates the image impedance \\\k\ the other of impedance Z2, the
image impedance W^k. The degree of simulation attained can be
seen from the characteristics of Fig. 16, wherein the effect of small
dissipation is included by assuming Vk = -\- -OlC/fc in a negative
branch and Vk = - .01 1/^ in a positive branch, as before. Over most
of a transmitting band the agreement is within a few per cent; outside
it is still quite satisfactory. Near the critical frequencies, where
" Still other forms of networks have been considered by R. Feldtkeller in a paper
"Uber einige Endnetzwerke von Kettenleitern," Eleklrische Nachnchlen-Techmk,
Band 4, Heft 6, p. 253, 1927.
322
BELL SYSTEM TECHNICAL JOURNAL
Uk = — 1. the simulation is improved by dissipation, as we might
expect.
This physical possibility of closely simulating the image impedance
of a wave-filter shows that the assumption of such a physical termi-
nation, as made in a previous paper,^^ was practically justified when
solving the problem of the behavior of wave-filters under non steady-
state conditions.
/
z
y
^>-
<^
V
/
"i
8
2"
\
^
.^
Real
/
/
y'
6
\
V
/
r
f
n
\^
/
1
~ A
(
1
2
2
L
;
2
1
'^r
2
7
I/,
-/.
2
J
-
2
z. l¥/me-Filterd,, y=^=^
y
~
a
^
^ —
~
6
-
8
Fig. 16— Simulation of the image impedances Wn and Wn- by the impedance net-
works of Fig. 15. (Broken lines are for dissipation with Vk = ± .01 Uh).
The particular structures for simulating the impedances of "con-
stant k" low pass, high pass, low-and-high pass and band pass wave-
filters, which correspond to the general ones of Fig. 15, are obtained
by terminating the networks of Appendix II with resistances R. It is
understood, of course, that others than the "constant k" wave-filter
of any class have either the image impedance Wik or W2k. Obviously,
it would be possible to simulate the impedance of any wave-filter which
by proper combination on the image basis can be linked with these
networks simulating PI'u or W^k- This, therefore, gives a method for
obtaining in a limited frequency range or ranges almost any resistance
characteristic with zero reactance.
Likewise, the impedance of a mid-series section of the shunt MM'-
type or a mid-shunt section of the series J/.l/'-type which has the
parameters of formula (32) and one pair of its terminals closed by a
1^ "Transient Oscillations in Electric Wave-Filters," J. R. Carson and O. J.
/obel. B. S. T. J., July. 1923.
ELECTRIC WA VE-FILTEK5
Hi
resistance R, is a good simulation of Wik{m, m') or W-iki'n, w')- The
latter are, as we know, approximately constant resistances equal to R
over desired frequency ranges and are reactances at other frequencies.
An interesting use of either or both of these simulating networks would
be as a balancing network against a resistance R or against each other
in a hybrid set. At frequencies in those ranges where the balance is
quite accurate, currents in the main circuit would be highly attenuated,
these attenuating bands corresponding to the transmitting bands of the
wave-tilter impedance section.
Part 4. Simulation of Loaded Line Impedances
The networks of Fig. 17 are capable of giving impedance simu-
lation over the greater part of the principal transmitting band of a
^>
which iwmuJbaJbeci' Kj
^
C3
o\\^
Z
J646L,j, .7250 Czh.
z-
I
-\$mj —
.2335 Ljh,
â– 5IJ0C2h
R
Z\
Jl4^u£-/i^cti/rn^ .inripecCamce /nehtrnvh
.luhidv Ainrrmlcube^ K^
Supplernentcuyy
— oAAAA-o-
Rjj C2,
HK^
Z.
B/iaic
.5J10L,^
.2,335 ^2;^ ^
3615 Czh
imczk
T
R
I
mmc^j.
Fig. 17 — Impedance networks which simulate the iterative impedances, Ki and K2, of
a loaded line at mid-load and mid-section terminations, respectively.
324 BELL SYSTEM TECHNICAL JOURNAL
loaded line. They are useful in cases where it is desirable to extend
nearer the critical frequency the range of simulation possible by
means of the networks described by R. S. Hoyt.'^
Designs are given for mid-load and mid-section terminations.
Results for other terminations can be obtained by building out the
load or section. From an economic standpoint it might be pointed out
that the basic networks for the mid-point impedances to be described
each have seven elements, whereas corresponding designs based upon
Figs. 14 and 15 of Hoyt's paper would have six elements. However,
the new mid-load basic network which extends the range of simulation
requires only one-half the total amount of capacity but slightly more
inductance than that required by the corresponding Hoyt network;
the new mid-section basic network requires only one-half the total
amount of inductance but slightly more capacity than the corre-
sponding Hoyt network.
4.1 Foundation of Designs
The design of any simulating network usually involves two processes,
namely, a determination first of structural form and second of mag-
nitudes.
The structural forms of the new designs follow readily from the well-
justified assumption that either mid-point impedance of a loaded line
in its principal transmitting band is approximately equal to the
corresponding mid-point impedance of a "constant k" low pass wave-
filter as the basic network, with the series addition of the impedance of
a supplementary network which simulates the additional impedance
introduced by dissipation at low frequencies. While this assumption
is really the same one which underlies the designs by Hoyt, the new
basic networks have considerably different forms and were derived
from wave-filter theory, which explains their inclusion in this paper.
In fact, the desired basic networks of Fig. 17 are immediately available
from the results of Part 3, being special cases of the networks of Fig.
15 which use the low pass wave-filters of Appendix H.
The particular supplementary network chosen, one already con-
sidered by Hoyt but designed differently, has four elements, two
resistances and two capacities, and is known to have the desired
impedance characteristic. The same one will generally do for either
mid-load or mid-section impedance, as it contributes impedance only
at the lower frequencies of the range.
The magnitudes of the elements of these networks are all determined
""Impedance of Loaded Lines, and Design of Simulating and Compensating
Networks," R. S. lioyt, B. S. T. J., July, 1924.
ELECTRIC WAVE-FILTERS 325
from computed loaded line impedances (or perhaps from measured
impedances), instead of directly from certain primary line and coil
data. This makes it comparatively easy to take account of variations
with frequency of the constants, such as line leakance and loading
coil resistance.
The mid-load iterative impedance is given by the formula
'^.^ -1- Sy\/ , , ^L „„,. ^^7
the mid-section iterative impedance by
'l+||coth^^
K, = k I ^ --' (39)
l+||tanh^
In these formulas y and k are the propagation constant and iterative
impedance, respectively, of the non-loaded line which may be computed
on the basis that the shunt capacity of each loading coil and its leads is
assumed to be concentrated, half at each end, and that each half is
added in the formulas to the line capacity of the adjacent section.
5' is the load spacing and Zl the load impedance.
4.2 Mid-Load Basic Netivork
This basic network has the structure and general design shown in the
upper part of Fig. 17. The magnitudes of its elements are fixed
when R and/o are known, since
Llk = R/T^fo,
and (40)
Cok = l/irfoR;
where R is the impedance ^LiklC-ik and /o is the critical frequency.
Its impedance in the frequency range considered is quite accurately
given by
which relation will be used for design purposes. The values of R and
/o are here determined for any particular loaded line by assuming that
at two frequencies, /a and/t, the corresponding values of r, respectively
r„ and r;,, are equal to the resistance components of Kx as computed at
those frequencies from (38). The frequencies /« and /& are chosen in
326 BELL SYSTEM TECHNICAL JOURNAL
the upper part of the desired range where the reactance components of
ivi are small. Substitution of these values in (41) gives two linear
equations in R~'^ and /o~- from which
. = . :'-^:^
'fa
and ^^' (42)
(1 _ ('-^
/r I \ fh'a
- Jl
l-(^^
The actual impedance, Z,, of the network with these values may be
computed as for any finite network.
4.3 Mid-Section Basic Nehvork
This network in the lower part of Fig. 17 is the mid-shunt simulating
network corresponding to Fig. 15.
Its impedance in the desired range is approximately given by the
formula
To determine R and/o, assume two values of r to be equal to Va and /-(,,
the resistance components of Ko as computed from (39) at two fre-
quencies fa and fb, where the reactance components of K^ are small.
Then from (43) we obtain two linear equations in R- and /o^- from
which
R
ifJ
<^-m
and (^-i)
The actual impedance of this network is Zo. The values of R and /o
from (44) will be practically the same as those from (42).
ELECTRIC WAVE-FILTERS 327
4.4 Supplementary Nehvork
Shown in both simulating networks of Im^. 17, this network has an
impedance expression of the form
flo + (i\if I • /A"\
\ + biif — hf-
where
a I ~ IttRiR^C-i,
b, = 27r{R,C2-\- R,C.2 + R,Q),
and
The resistance and capacity elements are obtained from the above
impedance coefficients as
Ri = aoa{'/(aQaibi — at^b-i — fli^),
Co = (flofli^i — 00^62 — fli-)/27rr/oT/i,
C3 = bojlirau (46)
and
Ri = oo.
From (45) the pair of impedance linear equations is
flo + fxbi + frbo = r,
and ^ (47)
/ai — //-&! -\- f'^xbi = X.
With the above formulas we can proceed to indicate the method of
design.
Ideally the network should have the impedance characteristic
z = r-h ix = K,- Zi, (48)
or
z = r + ix ^ K.- Z2, (49)
depending upon which mid-point impedance, Ki or K-z, is being simu-
lated. Usually these two values of z are practically the same. To fix
the four impedance coefficients, assume that the network has the ideal
components of (48) or (49) at two important low frequencies, the data
with increasing frequency being,
/i, rx + ixi ;
and
h, r^ + ix-i.
328 BELL SYSTEM TECHNICAL JOURNAL
These values are to be substituted in (47) to obtain four linear equa-
tions. The solution of these linear equations gives
Of) = ri — fiXibi — fi^fib-2,
«i = ribi — fiXib-2 + xi/fu
^^ fj^ihx, -hx,){r, - r.) + (.A-V2 -/â– a-0(/iVi -/2V2) ^5^^
, _ iMr, - r,)2 + (/,A-, - hx.Mf.x, - fuX2)
tH - j^ '
where
D =/./.{(/iVi -/.V2)(^i - r,) + (/ixi -f-2X.y}.
From the values of «o, fli, ^1. and b-. the network constants can be
computed by formulas (46). The network impedance is then given at
any frequency by formula (45).
The actual impedance simulating Ki is the sum, Z/ = Zi + s; that
simulating K2 is the sum, Z-/ = Z2 + -.
It should be pointed out here that the supplementary network may,
if desired, be given other structural forms having two resistances and
two capacities and having an equivalent impedance characteristic.
These other forms may be obtained by transformations from the
known one above or their elements determined from other formulas
corresponding to those of (46).
Likewise, a supplementary network which has a smaller or larger
number of elements than the one above might be used satisfactorily
with the same basic networks or their equivalents. That depends upon
the low-frequency impedance characteristics of the given loaded line
and upon the closeness of simulation desired.
4.5 Application of Results
To illustrate the possibilities of these impedance networks, mid-load
and mid-section designs are given here for a 19-gauge B-88-50 loaded
side-circuit. The "5 " spacing is 5 = .568 mile (3000 feet).
Data for the mid-load basic network, taken from computations of
A'l, are
fa = 3000, I'a - 1324;
and
fb = 5000, n = 720.
These give from (42), R = 1564.4 ohms, and /o = 5632 cycles per
second.
Data for the mid -section basic network, taken from computations
ELECTRIC WAVE-FILTERS 329
of A'^ are
fa = 3000, ra = 1848;
and
/ft = 5000, n = 3387.
Then from (44), R = 1564.6 ohms, and /o = 5638 cycles per second.
Because of the close agreement between these two sets of results,
their approximate mean values will here be used in both basic networks,
namely
R = 1565 ohms,
and
/o = 5635 cycles per second.
With these values in (40), Ln, = 88.38 mh., and Cok = .03611 mf. We
have then for the mid-load basic netn'ork the inductance and capacity
elements:
.3615 Ln = 31.95 mh.; .2335 L^ = 20.64 mh.;
.1646 Lifc = 14.55 mh.; .1494 L^ = 13.20 mh.;
.5110 Cat = .01845 mf.; .7250 Cat = .02618 mf.;
and for the mid-section basic network
.5110 Lu- = 45.16 mh.; .7250 Lu- = 64.08 mh.;
.3615 Cot = .01305 mf.; .2335 Cu = .008431 mf.;
.1646 Cofc = .005943 mf.; .1494 du = .005395 mf.;
wath their locations as in Fig. 17.
The impedance characteristics of these basic networks, Zi and Z-2,
were computed directly from the finite networks on the assumption of
small coil and condenser dissipation constants, d = d' = .005. Com-
paratively small reactance components begin to appear above 4500
cycles per second. Increasing the amount of dissipation in the
reactance elements would tend to increase the reactance components of
Zi and Zo at the upper frequencies.
The design of the single supplementary network w^as made from low
frequency data representing the average values of {Ki — Zi) and
(ivo — Zo). The data are
/i = 100, ri + ixi = 152 - ilOO,
and
/2 = 300, r. + ix2 = 20 - i252.
From formulas (50) we obtain
flo = 7839.0; ai = 233.12;
bi. = 17.600-10--; b-z = 30.481 -10-*.
330
BELL SYSTEM TECHNICAL JOURNAL
From (46) these give
Ri = 5327 ohms;
Q = 2.081 mf.;
G = .8886 mf.;
R. = 7839 ohms.
The impedance characteristic above 100 cycles per second as computed
from formula (45) is mostly that of negative reactance, both com-
ponents decreasing rapidly with frequency.
6000
.5000
4000
3000
^
zooo
1000
AT^t „^; J ^- Mid-.hoQxL, Z,
\2, Mi/jb-Cvectlcrry, Zz
/
)
f
/
^
^
,
n&iicxla/nce
^^
â– ^
â– N,
\
I ,
r
^ lb
00
20
00
"icy
30
oy-clec
00
y -per
40
00
rUjC)
30
^
1
Reaa
lam<je
2
-/ooo
Fig. 18 — Simulation of the iterative impedances, Ki and K2, of a 19-Ga. B-88-50
loaded side-circuit by the impedance networks of Fig. 17. (Coil and condenser
dissipation constants are d = d' = .005.)
Final results showing the characteristics of the complete simulating
networks are compared with those of the loaded line in Fig. 18.
ELECTRIC WAVE-FILTERS 331
Simulation is within .7 per cent of the impedance over the continuous
range from 100 to 3000, within 2 per cent from 3000 to 5000, and within
4 per cent from 5000 to 5500 cycles per second; the per cent accuracy
is best in the case of the mid-section network. This upper frequency is
approximately 97 per cent of the critical frequency, 5635 cycles per
second. There is good simulation even considerably beyond the
critical frequency, as may be inferred from Fig. 16.
For still greater precision, networks which originally have three or
more parameters and which are formed in a manner similar to those
of Fig. 15 may constitute the basic networks.
4.6 Other Approximate Designs
Alternative designs of networks simulating Ki and K-i can be made
with the networks of Fig. 15 as foundations. The method of doing
this will merely be outlined here since the networks do not appear to be
as practical as the ones already described in detail.
This procedure assumes that the actual loaded line structure can be
quite accurately represented physically in the desired frequency range
by a ladder structure of series and shunt impedances, Zi and Zi, re-
spectively. Roughly, Zi would be series resistance and inductance and
So would be parallel resistance and capacity. Then throughout the
two networks of Fig. 15 the impedance of Zik is to be replaced by that
of 2i and the impedance of z-ik by that of z-z- Also the terminating
resistance R is to be replaced by V21Z2, the impedance of the corre-
sponding uniform line, which in this case might be approximately
simulated by a resistance in series with a network like the supple-
mentary network of Fig. 17. The resulting impedance networks
would then approximately represent Ki and K^. However, no design
formulas are needed to show that even if these networks give as good
simulation as the networks of Fig. 17 they would require more elements.
Appendix I
Reactance Frequency Theorems and Proofs of Frequency Relations in
M-Type or Higher Order Wave- Filters
There are certain simple frequency relations which hold in the
reactance characteristics of non-dissipative impedances. A statement
and proof of these relations will first be given. From them will follow
readily the proofs of the frequency relations in the characteristics of
M-type or higher order wave-filters, which are represented by formulas
(20) to (24), since they require a consideration of the "constant ^"
series impedance Su only.
332
BELL SYSTEM TECHNICAL JOURNAL
Reactive Impedance Characteristics
All non-dissipative impedances have reactances which can be
separated into four forms of impedance functions, each of which can be
expressed as the ratio of two frequency-polynomials in if, where
i = V— 1, and / is frequency. It is known that such a reactance
necessarily has a positive slope with frequency and hence the resonant
and anti-resonant frequencies alternate on the frequency scale. The
four mathematical forms may be separated on the basis of the general
location of their resonant frequencies and have finite resonant fre-
quencies with or without zero and infinite resonant frequencies. These
reactive impedance forms are as follows:
Form 1. Resonant at zero and w finite frequencies.
_ a,if + a.iify + ■■■-f fl,„+,(-//)^»
1 + b.xifr + • • • + bu^ir-"
+1
= tx.
Form 2. Resonant at n finite and infinite frequencies
1 + aodfY- + h a^ndfy"
= tx.
bvlf + b,{ify + • • • + 62n+l(//)^"+^
Form 3. Resonant at zero, n finite and infinite frequencies.
_ a,if+aSff + h n2n+x{iff"+'
^ 1 + b.iiff + • • • + 62n+2(^/)2"+2
Form 4. Resonant at n finite frequencies.
1 + aSfY + • • • + a2n((f)-"
= IX.
z =
b^f+b^{ifY+ ••• + b,n-,iiff--'
(51)
(52)
(53)
(54)
Each of these forms has a simple frequency relation which is expressible
as a theorem.
Reactance Frequency Theorems
The product F of the frequencies at which the reactance x is ± c in each
of the four reactive impedance forms is the folloiving:
Form 1.
Form 2.
Form 3.
F'^n+i = , proportional to c.
a^n+i
F2n+i = -T . inversely proportional to c.
COoji+i
F-2n+2 = -J , independent of c.
0-271+2
ELECTRIC WAVE-FILTERS 333
When r = <x , meaning anti-resonance of z, each anti-resonant frequency
appears twice in the product.
Form 4. F^n = — , independent of c.
fl2n
When c = 0, meaniyig resonance of z, each resonant frequency appears
twice in the product.
To prove the theorem for Form 1 first square the expression in (51)
and clear the fraction. This gives a polynomial in /^ of degree
2w + 1, of which only the terms of highest and zero powers need be
shown for our purpose. Thus
(j2)2„+i _,_ -1^ = 0, (55)
which expresses the general relationship between x^ and /-. If x^ is
given some constant value as x^ = c^, that is x = ± c, the roots of (55)
will be the 2w + 1 distinct values of /^ where x = ± c. By the theory
of equations, the product of these 2n -\- I values of /^ is (c^/aon+i).
Since we are interested only in positive frequencies, we may take the
positive square root of both sides with the result that the product of all
frequencies at which x = ± c is c/a2n+u which proves the theorem.
The proofs of the theorems for Forms 2, 3 and 4 are exactly similar
and should not need further explanation. In Form 3 the values
.X- = -£- GO occur at the anti-resonant frequencies of s, namely fia, f^a,
etc. ; hence, when c = oo the total frequency product includes each of
the latter frequencies twice. The result for Form 4 has a meaning
even at the limit c = 0. These frequencies are the resonant ones of s,
where z = 0, and each one of them must obviously appear twice in the
total product.
Proofs of Wave- Filter Frequency Relations
As was stated in Section 1.9, Zik satisfies certain conditions at tiie
particular frequencies of interest.
At critical frequencies, /o, /i, etc.,
zik = ± i2R. (56)
At frequencies of infinite attenuation, /o^), /ico. etc.,
zik = =L (o7)
V 1 - 2-
334 BELL SYSTEM TJiCHNICAL JOURNAL
l-'very negative or positive branch of Zu: includes one each of these
frequencies.
For those wave-filters with only internal transmitting bands the
additional relation will be used which specifies the frequencies where all
image impedances equal R and the series impedances become resonant.
At these resonant frequencies, fir, fir, etc., in the transmitting bands
zik = 0. (58)
We know that in a "constant ^" wave-filter the transmitting bands
include the frequencies at which the series impedance Su is resonant.
Hence, to the four forms of impedance function for Ziu, as in (51) to (54),
there correspond four groups of wave-filter classes as already men-
tioned. These groups were designated according to the general
locations of their transmitting bands which obviously correspond to the
locations of the resonant frequencies of su-. For this reason each wave-
filter group and the corresponding impedance form of Su have the same
number designation.
Group 1. Low-and-w Band Pass.
An application of the theorem for Form 1 with (56) and (57) gives
immediately the desired relation (20)
./Ooo/loo • • • /2nco = f ^ „ -•/"./> ' ■* /2»-
VI - g"
Similarly the relations (21), (22) and (23) are obtained for Groups 2,
3 and 4. Relation (24) for Group 4 is derived from (56) and (58), the
latter corresponding to c = in the theorem for Form 4 where each
resonant frequency appears twice; the square root of the resulting
relation is (24).
ELECTRIC WA VE- FILTERS
335
Appendix II
Fixed Terminal Transducers of Several Wave-Filter Classes
I. Low Pass.
.1646 Ljh y^50 Czh
.3615 L,K
-o o '^'WS>>-
<K^mj--
w., w.
% '^rk
^ O-
I
5JWC2K
W;^,(7rk?rh) %R
-o o-
U.-Shnml lemnimnl MxjmcujCacer
.5110 L,h
-o o-
.Z5d)5 Czh,
-o o-
^1646 CzK
-o o-
-o o-
r - ^ r ^
336
BELL SYSTEM TECHNICAL JOURNAL
II. High Pass.
llj-S^rJ^eciy JjemrwnoJb Mxi/ru^ducer
1.379 Iz^, 6.076 C,h,
Z.766C,ji r-<^5W>>-HH— 1
-o o Hh f '^
—HI-
6691 C,h
^, H,
-o o-
I
l957L2h
aZ8Z C,h
I%/^fm,7n;) </v
-o o-
JlzrSMiml Mfymmoub 2fxmAducer
4.28ZL2h,
(i •
-o o-
l957C,i^
-o o-
/r* ^*
-o o-
^ \l373Cjj,
ka69/I.zh
-o o-
Cu: =
1
47r/ii? '
Lo,
R
47r/i
ELECTRIC WA VE-FILTERS
337
III. Low-and-High Pass.
111,7 Sj^^ie^ J^y?ni/nal/ JjzamAdhcer
.36J5Ljh
(I • ty
â– â– Hk ^.
23351,,^ C^
v-^W^ /.379L2j^
.1— <r5w>' — o-|[-o — ,^-o
HM .7Z50C2},
6.076C,h
Hh-
1
l957L^j^
5110 C^h
6.691 C,f,
IIlzrSTia/nl ]:£r/mifnali Ira/riMiUyOer
..5//0L,j^
K, C
ilf|^^f£^ .
I Z. 766 -L^k
-r..36/5C2h
/.957C,f,
6.076 L^T^
\~\379Cjh,
7Z50L,j,\
T
K.
'2h('rrh,m)\
6.69JLzj,\
/6a6Czj^
.1494 qj.
.R
Cik =
1
47r(/i - f,)R '
L
R
'' 47r(/i - /„) '
/i"/o
TrfofiR
338
HELL SYSTEM TECHNICAL JOURNAL
I V. Band Pass.
IV,.- S^ae^ 2epmmcLb Jrwrmdjuuoer
.1646 L,h a076C,H, KWH
I — 'f^W^ — HK-^'
. 36/5 Lfji, 2. 766 Cjh
o — rfijb^P^ — HK
«^*
-<LMib-
7Z60C2K
<>— o
I.957L
'Zhj<-
1
J"-
SlWCzh
.2335 L,f^ g MmLih, 6.69J Cih
a.282C>h W,^7rv,7rv') tlb
l]/2~SM(/nb Jery??iimal/ Jjxi/nddzocer
2.7661.2b
.5/I0L;h W57C,h
^<M^ — HI-*— ^
7250 L,K
t^lt — ~77 TT » /^2A- : — ■:r-.
C\]c —
h - /l
C,
47r/,/2 '
1
t(/2 - /l)i?
ELECTRIC WA VE-FILTERS
339
V. Equivalents of Fixed Terminal Transducers of Fig. 11.
V,-
O o-^VVW' —
.5615 Z;/^
m
ih
1.078 Z,h% .398JZ,h
4.734 ZzH, I .^•^^^■Z^;^:
l%/f^(7n,'rrh)
^zr .21IZZ,K .2998 Z,f,
a-
W,^ %6.691Z^^
9Z78Zz}^ I Z.SIZZzfy
Z. 766 Zzh W2X, (rrv, rri)
— o
340
BELL SYSTEM TECHNICAL JOURNAL
Appendix III
Equivalent Transducers and Transformation Formulas
Transformation 1
4 o
-o-VSAA-'*-
Equivalent when
b = a(\ + a), c = 1 + a.
Transformation 2
Equivalent when
b =
c =
1 + a
16 For transformations of simple equivalent two-terminal or impedance networks
containing two kinds of general impedances, see Appendix 111 of paper in footnote 1.
Also U. S. Patent No. 1,644,004 to O. J. Zobel, dated October 4, 1927.
Appendix IV
341
1
1 ^
\
«t
\
1
\\
'^
\
\
to
1
N
§
o
-^
'
00
1
<yi
O
1
1
^
u
g^
(^-W&Ofe^^J/j
-^
Abstracts of Technical Articles From Bell System Sources.
Western Electric Remodels Power Plant at Ilaivthorne Works} C. B.
Barnes. The summary of a six-year revamping program. A fea-
ture of the new plant is the installation of the largest cooling towers in
America. Airplane propeller-type forced-draft fans are employed.
Long Telephone Lines in Canada? J. L. Clarke. This paper de-
scribes the development of the long distance telephone service in
Canada, historically, from its inception and the installation of the
nucleus of 360 miles, up to and through the present status and lines
listed in Table I, to the proposed development represented by Table II,
the result of a careful study of calls per day to be expected by 1932.
This effort is to provide for traffic requirements in a manner most suit-
able from a transmission point of view, and to accomplish it with a
minimum amount of switching. Much of the engineering work for
this is already actively under way and certain work of construction
actually commenced. A survey of existing routes and the matter of
transmission maintenance are discussed.
The ''Raman Effect.'' ^ C. J. Davisson. A brief and informatory
account of the " Raman Effect." For this new discovery in the realm
of light and spectra, appraised as one of the most important achieve-
ments in physics in recent years, Sir C. V. Raman of India was awarded
the Nobel Prize in physics for 1930.
Planning a Plant for the Manufacture of Lead-Covered Telephone
Cable} J. C. Hanley. Results of a study to determine the size and
type of building to be erected, the arrangement of machinery for the
most direct handling of product during process of manufacture, and the
most efficient materials-handling equipment.
Outdoor Atmospheric Corrosio7i of Zinc and Cadmium Electrodeposited
Coatings on Iron and Steel.'' C. L. Hippensteel and C. \V. Borgmanx.
Experimental data are presented on the rates of corrosion of electro-
plated zinc, zinc alloy and cadmium protective coatings on steel in a
1 Power, Dec. 2, 1930.
•' Jotir. A. I. E. E., Dec, 1930.
^Sci. Monthly, March, 1931.
' Mech. Engg., March, 1931.
* Trans. Atner. Electrochemical Soc, \'ol. I.\ III, 1930.
342
ABSTRACTS OF TECIIXICAL ARTICLES 343
severely industrial atmosphere, and in a similar atmosphere, but ac-
celerated by additional rainfall simulated by a water spray. These
data show that zinc and zinc alloy coatings corrode at a slower rate than
cadmium coatings. However, under the accelerated exposure the
difference is not so pronounced.
Telei'isiov in Color from Motion Picture Film.^ Herbert E. Ives.
In speculations on the possible uses for television, one project which
receives considerable attention, partly because of its relative ease of
accomplishment, is the transmission of images from motion picture
film. It is true that the practical simultaneity of event and viewing,
which is the unique offering of television, is lost when the time necessary
for photographic development of the film intervenes. Nevertheless,
it is conceivable that if this delay is small, television from film may still
possess such an advantage over the material transportation of film as to
give it a real field. A further possibility, more remote, but within the
range of legitimate speculation, is that television apparatus may some-
time be used to receive, in the home, motion pictures of the sort now
offered in the theatres or in home projection outfits. Howe\-er distant
these mergings of the two arts may be, the technical problems presented
are pretty clearly defined, and offer interesting features for study.
Among these problems, and perhaps the farthest cry of any, is the
transmission of images in color from colored motion picture film. This
paper describes a method of accomplishing this, using the receiving
apparatus for television in color recently described, and special sending
apparatus which utilizes the latest form of colored moving pictures
— the ridged film now marketed under the name of Kodacolor.
Private-Wire Telegraph Service.' R. E. Pierce. An important
part of the entire communication service of the United States is de-
voted to private wire service. More than one and one-half million
miles of private wire telegraph service is furnished to press associations,
brokers, financial houses, public service companies, and other organiza-
tions and individuals. Some of the interesting features involved are
described here.
Absolute Amplitudes and Spectra of Certain Musical Instruments and
Orchestras.^ L. J. Sivian, H. K, Dunn, and S. D. White. In a
paper on "Speech Power and its Measurement," one of the authors
has given some measurements of average and peak amplitude in speech,
6 Jour. Op. Soc. Amer., Jan., 1931.
-I Elec. Ewgg., Jan., 1931.
» Jour. Acous. Soc. Amer., Jan., 1931.
344 BELL SYSTEM TECHNICAL JOURNAL
using apparatus in which the speech spectrum was divided into thirteen
bands of frequencies. The same apparatus has been used in a series of
measurements on musical instruments, which are reported in this paper.
As with the speech measurements, the data are statistical in nature,
and are taken with a view to their engineering applications. These
applications are concerned, chiefly, with the transmission and reproduc-
tion of music, and the data should show the power and frequency re-
quirements for systems which are called upon to perform these func-
tions without distortion. In carrying out this purpose it has been
thought well to measure both individual instruments, and instruments
playing together in orchestras; to make measurements on actual musi-
cal selections, rather than on single notes; and to take the measure-
ments in such a way as to obtain an average or integrated picture of the
selection, as well as the distribution of amplitudes in magnitude and
frequency, the extreme values being particularly important.
Noise Measurements.'^ John C. Steinberg. That noises have a
detrimental effect upon human health and happiness has been proved
and now efforts are under way to control or eliminate objectionable
sounds. Some of the problems involved are outlined and a newly
developed "noise meter" is described.
Fatigue Studies of Telephone Cable Sheath Alloys}^ J. R. Townsend
and C. H. Greenall. This paper is a continuation of a previous pa-
per presented before the Society by one of the authors in 1927 and
further discusses results of fatigue studies of lead sheath for telephone
cables. The results of the investigation of the fatigue characteristics
of lead cable sheath alloys, using the rotating-beam type fatigue
machine, are reported. Data are also given for static fatigue.
The failure of lead cable sheath alloys as reported in the previous
paper is by intergranular fracture and in the case of the lead-antimony
alloys repeated stress appears to reduce the solubility of antimony in
lead. The type of fracture observed for the rotating beam speci-
mens is similar to that of the repeated flexure specimens described in
the previous paper. The type of failure on the static fatigue test is a
breaking down of the bond between the crystals.
The fatigue properties of the 0.04-per cent calcium-lead alloy de-
scribed in this paper are by intergranular fracture, but there is no loss
of solid solubility of the calcium in the lead. Great improvement in
^ Elec.Engg.,ydi\., 1931.
^^ Proc. Amer. Soc.for Testing Materials, \'ol. 30, Part II, 193U.
ABSTRACTS OF TECHNICAL ARTICLES 345
the fatigue endurance was noted for an alloy of the same tensile prop-
erties as the lead-antimony alloy.
A Cooperative Electrolysis Survey in Louisville, Kentucky.^ W. C.
White. A cooperative electrolysis survey in the city of Louisville,
Kentucky, under the direction of an electrolysis committee is described.
An analysis of a portion of the survey data and indicated mitigation
measures are given as typical examples. The advantages of coopera-
tive action in a general electrolysis survey are shown.
^Elec. Engg., Feb., 1931.
Contributors to this Issue
O. B. Blackwell, B.S. in electrical engineering, Massachusetts
Institute of Technology. After graduation, he entered the Engineering
Department of the American Telephone and Telegraph Company and
in 1919 was made Transmission Development Engineer. Mr. Black-
well has general supervision of transmission developments and has
been prominently associated with progress in long distance wire and
radio telephony.
R, N. CoNWELL. Mr. Conwell is Transmission and Substation
Engineer, Public Service Electric and Gas Company, Newark, New
Jersey.
Lloyd Espenschied. Mr. Espenschied is High Frequency Trans-
mission Engineer, Department of Development and Research, Ameri-
can Telephone and Telegraph Company. He joined the Bell System
in 1910, having graduated from Pratt Institute the previous year.
He has taken an important part in practically all of the Bell System
radio developments, beginning with the first long-distance radio-tele-
phone tests of 1915, at which time he received the voice in Hawaii from
Arlington, Va. He has participated in a number of international
conferences on electric communications.
Bancroft Gherardi, B.Sc, Polytechnic Institute, Brooklyn, N. Y.,
1891; M. E., Cornell University, 1893; M.M.E., Cornell University,
1894. New York Telephone Company, Engineering Assistant, 1895-
99; Traffic Engineer, 1899-1900. New York and New Jersey Tele-
phone Company, Chief Engineer, 1900-06. New York Telephone
Company, and New York and New Jersey Telephone Company,
Assistant Chief Engineer, 1906-07. American Telephone and Tele-
graph Company, Equipment Engineer, 1907-09; Engineer of Plant,
1909-18; Acting Chief Engineer, 1918-19; Chief Engineer, 1919-20;
Vice President and Chief Engineer, 1920-. Mr. Gherardi is a Past
President of the American Institute of Electrical Engineers and is now
President of the American Standards Association.
William H. Harrison, Plant Engineer, American Telephone and
Telegraph Company. Mr. Harrison entered the Bell System in 1909
as a repairman for the New York Telephone Company. In 1915 he
became engaged in circuit design work with the Western Electric Com-
346
COXTRIBUTORS TO THIS ISSUE 347
pany and in 1918 joined the staff of the American Telephone and
Telegraph Company. He was made Equipment and Building Kngi-
neer in 1924, Acting Plant Engineer in 1928 and Plant Engineer in 1929.
H. L. HuRER, Cornell University, 1909-13; Chesapeake and Poto-
mac Telephone Company and Associated Companies, 1913-17;
Signal Corps, U. S. Army, 1917-19; Chesapeake and Potomac Tele-
phone Company and Associated Companies. 1919-27; American Tele-
phone and Telegraph Company, Department of Operation and Engi-
neering, 1927-. Mr. Huber is now Engineer on Foreign Wire Relations.
Herbert E. Ives, B.S., University of Pennsylvania, 1905; Ph.D.,
Johns Hopkins, 1908; assistant and assistant physicist, Bureau of
Standards, 1908-09; physicist, Nela Research Laboratory, Cleveland,
1909-12 ; physicist, United Gas Improvement Company, Philadelphia,
1912-18; U. S. i\rmy Air Service, 1918-19; Western Electric Company
and Bell Telephone Laboratories, 1919 to date. As Director of Elec-
tro-Optical Research, Dr. Ives has to do principally with the produc-
tion, measurement and utilization of light in communication problems.
J. C. Martin. Mr. Martin is associated with the Middle West
Utilities Company, Chicago, Illinois.
Edward C. Molina, Engineering Department of the American
Telephone and Telegraph Company, 1901-19, as engineering assistant;
transferred to the Circuits Design Department to work on machine
switching systems, 1905; Department of Development and Research,
1919-. Mr. Molina has made contributions to the theory of proba-
bility and its applications to telephone problems, such as the efficiency
of various trunking arrangements and the significance of data derived
from samples. He has also taken out several important patents re-
lating to machine switching.
R. F. Pack. Mr. Pack is Vice President and General Manager,
Northern States Power Company, Minneapolis, Minnesota.
A. E. Silver. Mr. Silver is Consulting Electrical Engineer, Electric
Bond and Share Company, New York, N. Y.
H- S. Warren, A.B., Stanford University, 1898. American Bell
Telephone Company 1899-1903; American Telephone and Telegraph
Company, 1902-. Department of Development and Research, 1919
to date; now Protection Development Engineer. Mr. Warren's work
has been chiefly of a development character in the field of transmission,
equipment, and electrical interference.
348 BELL SYSTEM TECHNICAL JOURNAL
H. L. Wills. Mr. Wills is Assistant to X'ice President and General
Manager, Georgia Power Company, Atlanta, Georgia.
WiLLLVM Wilson, Victoria University of Manchester, 1904-10;
B.Sc, 1907; M.Sc, 1908; Cavendish Laboratory, Cambridge Univer-
sity, 1910-12, B.A., 1912; Lecturer in Physics, Toronto University,
1912-14; D.Sc. Manchester, 1913. Engineering Department, Western
Electric Company, 1914-24; 1925- Bell Telephone Laboratories; Assist-
ant Director of Research 1928-. Dr. Wilson has published numerous
papers on radioactivity and thermionics and since 1917 has been in
direct charge of vacuum tube development and design and since 1925
has also been in charge of radio development.
O. J. ZoBEL, A.B., Ripon College, 1909; A.M., Wisconsin, 1910;
Ph.D., 1914; instructor in physics, 1910-15; instructor in physics,
Minnesota, 1915-16; Engineering Department, American Telephone
and Telegraph Company, 1916-19; Department of Development and
Research, 1919-. Mr. Zobel has made important contributions to
electric circuit theory, which includes the subject of distortion correc-
tion as well as that of wave-filters.
The Bell System Technical Journal
July, 1931
Some Physical Characteristics of Speech and Music *
By HARVEY FLETCHER
Kinematic and statistical descriptions of the physical aspects of speech
and music are given in this paper. As the speech or music proceeds, the
kinematic description consists in giving the principal melodic stream,
namely, the pitch variation and also the intensity and the quality variations.
For speech and song, the quality changes are principally described by giving,
besides the main melodic stream, two secondary melodic streams correspond-
ing, respectively, to the resonant pitches of the throat and mouth cavities.
To this must also be added the positions of the stops and the high pitched
components of the fricative consonant sounds as functions of the time. The
statistical description consists in giving the average, the peak, and the
probable variations of the power involved as the various kinds of speech and
music proceed. These general ideas are illustrated by numerous experi-
mental data taken by various instrumental devices which have been evolved
in the Laboratories during the past fifteen years.
A speech or musical sound is transmitted from the mouth of a speaker
or from a musical instrument through the air to the ear of the
listener by means of a pressure wave, a succession of condensations
and rarefactions of the air. Such a wave spreads in all directions
away from the source of sound and soon encounters solid objects which
cause reflections. These reflected waves combine with the original
one and thus modify the pressure changes taking place at any point.
In this paper we shall be concerned chiefly with the pressure changes
which take place before reflections occur.
Speech is composed of fundamental sounds called vowels and
consonants. As a conversation proceeds there is a constant shifting
from one of these sounds to another, only one of them being sounded
at one time. Most of these sounds may be continued as a steady
tone and hence may be designated as continuants. The others require
that the sound stream be interrupted and are therefore called stops.
The first class includes the long and short vowels, the diphthongs, the
semi-vowels, and the fricative consonants, the sounds a, i, ou, 1 and s
being typical, respectively, of each of these groups. The pure stops
are p, t, ch, and k. In producing the corresponding voiced stops,
b, d, j and g, the voiced stream is not entirely interrupted, although
the tones from the vocal cord are very much subdued. A conversation,
* Presented as invited paper in Symposium on Acoustics, American Phys. Soc,
Dec. 30-31, 1930, Cleveland, Ohio. Published in Rev. of Modern Physics, April, 1931.
349
350
BELL SYSTEM TECHNICAL JOURNAL
then, consists of a succession of continuants and stops and a physical
interpretation of speech consists, therefore, of a description of these
continuants and a discussion of the manner of joining the continuants
together either directly or by means of stops.
Melodic Streams of Speech
As an example of how this analysis of speech may be made consider
the sentence, "Joe took father's shoe bench out," an oscillogram of
which is shown in Fig. 1.^ This silly sentence was chosen because it
.J I Oi ' ' ' ' ' ' i
â– ' 1 1 I 1 I I ' 2
f â– I â– r - I
R
( 6
2 7 SH
'\-'-'
.^.•H'V/V,'\/v'V^yV^/.A,,AA/^^/
V^A-,^■'-:V''A^■.,V-'K '.•\''-X"/-^"'x-."/-'^^-- '.■••'
I I
Fig. 1— Oscillogram: "Joe took Father's shoe bench out" — spoken.
is used in our laboratory for making tests on the efficiency of telephone
transmitters. This sentence together with its mate "She was waiting
at my lawn" contains all of the fundamental sounds in the English
iThis oscillogram and the others following it were taken with the new high
quality and high speed oscillograph which has recently been developed m our labora-
tory. It has an approximately uniform response for amplitude and phase from 20
to 10,000 cycles per second.
PHYSICAL CHARACTERISTICS OF SPEECH AND MUSIC 351
language that contribute toward the loudness of speech. In Fig. 1
the ordinates are proportional to the pressure change in bars and the
abscissas are time intervals of .01 second. The eighteen fundamental
sounds in this sentence are joined together without the stream of sound
being interrupted except for the stops t, k and ch. The stop consonant
b is voiced so that although the vocal cord sound is interrupted by
the closing of the lips, it continues to sound in a subdued way until
the stop is removed and the e sound begins. Pauses, that is, silent
SILENT
-*1
r ^
l-SILENf
-1 t*SILENT
3
1
1
z
li
Ch
i
1
L
nr
sh
' i
b
t
It
"^w
/
^
to
1^
y^
^^_
7
/
/
/
A
\
> ^
6
i
^1
V
1 ^
^
4
1
7
'^
O
z
J
3 1
1 3
\
\
\
/
^
1
I
u
1-
f\
V
1
\ 3
»
2
/
^<
'<
Q.
-2
^
L
J
r
'v.
t
1 ^
•
N
V
-3
N
^ 1
^
J
t
u
K
f a
th r z
sh u b
en c
h a u t
1
f
\
7
5
IME
N SE
3
CONC
1
)S
1
2
1
,4
Fig. 2 — Melodic curves: "Joe took Father's shoe bench out" — spoken.
intervals, are made between sentences and sometimes between words.
It will be noticed that a brief pause was inserted at the intervals .17
to .21 and .32 to .335 and .34 to .41 and 1.16 to 1.18 seconds. There is
no such pause between "shoe" and "bench."
Speech, then, consists of a series of comparatively steady states of
vibration joined together in time, either by silences or transitions from
one steady state to another. Each one of these steady states is
characterized by a pitch and a tone quality, and the sequence is
352 BELL SYSTEM TECHNICAL JOURNAL
essentially a melody. The melody of the sentence whose wave form
is shown in Fig. 1 may be illustrated graphically as indicated in Fig. 2.
In this figure the ordinates represent the pitch in octaves below or
above a tone having a frequency of one kilocycle per second; or if
the frequency / is measured in kilocycles, then the pitch P is given by
the equation
P = \0g2f. (1)
The abscissas represent the time in seconds. The lower curve gives
the changes in the pitch of the fundamental and represents the melody
as ordinarily understood in music. The middle two curves represent
the pitch positions of the strongest harmonics. The location of these
positions is determined by the resonant properties of the throat and
mouth cavities. These curves may be considered as secondary melodic
streams. The combination of these two secondary melodic streams is
interpreted by the senses as a sequence of spoken vowels rather than
as a series of pitch changes. The small number above each part of
the curve gives the number of the harmonic which is augmented by
the resonance of the mouth or throat. For the sound e in bench the
4th harmonic was the strongest at the beginning of the sound, but
the 5th came in strongest near its end. I have tried to indicate the
relative intensities of the harmonics as the sound proceeds by the rela-
tive thicknesses of the lines. An examination of the oscillogram shows
that the intensity of the harmonic always increases as its pitch becomes
nearer the characteristic pitch for the vowel being spoken.
As indicated by the short lines at the top of the chart, there exists
at certain intervals high pitched components which are characteristic
of the fricative sounds. The unvoiced sounds t, k, f, z and sh, exist
only when the three melodic streams are stopped. The high pitched
components of the voiced sounds, j, th and b, are superimposed upon
the three melodic streams.
Besides these four important streams of speech (Fig. 2), there are
a great many others with intensities which are in general much lower,
but when combined with the main streams they determine the kind
of voice, that is, whether it is smooth and musical or rough and
harsh. The main melodic stream for a woman's voice is between the
pitches — 1 and — 2 octaves while for a man's voice it is between
— 3 and — 2 octaves. The secondary melodic streams produced
while speaking the same sentence are approximately the same for man
and woman and of pitches shown in Fig. 2.
In Fig. 3 is shown an oscillograph of the sentence "How are you?".
PHYSICAL CHARACTERISTICS OF SPEECH AND MUSIC 353
This sentence contains no stops. The sound stream is not interrupted ;
it is just a continuous variation from one vowel to another. In Fig. 4
the main melodic stream is given.
'
Fig. 3 — Oscillogram: "How are you?"
In Fig. 5 an oscillograph of the sentence "Joe took father's shoe
bench out" is shown when the vowels of this sentence are intoned on
the simple melody do-re-me-fa-me-re-do, and in Fig. 6 the melodic
a
u
'
^
—
^^
+-•â–
1
S--
_5(y^
4-,
u
4
"^
TIME IN SECONDS
Fig. 4 — Melodic curve: "How are you?"
streams are given. In this case only the characteristic resonant
pitch positions for the two secondary melodic streams are given. The
chief difference between this figure and that for the spoken sentence is
354
BELL SYSTEM ^TECHNICAL JOURNAL
in the main melodic stream. For purposes of comparison the curves
of the spoken and sung sentence are enlarged and shown together in
Fig. 7. In the case of the sung sentence the pitch changes are in
definite intervals on the musical scale while for the spoken sentence
o
;*A ,'\>t^'\''-'Vv.*
;vx-//-v~-//-^"-.-.^^-w,\>.-v^\-v.'v^v
I III
'- ' , -~^^C â– ' ^^^'^ '
I I
./'â– â– \-'/^ .;:\-'.-'^
•F
' SH ' lb
■^<'; -.'V'*'^' -.'Vvv' •y%''^'V' -''-''^^ -.-'^''^^ vvV^-
CH
U 17
. TH I 8 I
^1 I I
/x-^'-^'^V'^v-^V'^'^''vV'^
' I I
â– â– A.>>.'
>^-'
."â– A.-^^Vs-'vv^
-^7
u
I
13
."vV"^
1 I IT I ' ' â– ' '19
Fig. 5 — Oscillogram: "Joe took Father's shoe bench out" — sung.
the pitch varies irregularly, depending upon the emphasis given.
The pitch of the fricative and stop consonants is ignored in the musical
score, and since these consonants form no part of the music they are
generally slid over, making it difficult for a listener to understand the
PHYSICAL CHARACTERISTICS OF SPEECH AND MUSIC 355
3
5
a'
1
J_
ii
_sh
b
2
(
±_
/"
"—
"^^
lO
,
— .
1
'â– ^
^
>
<
o
- "
â– ''
^-
/
,,-—
-'-
-
o
/
^^
r
I
/
>
— .^
-—
''
o -1
"^
'
â– v^
,
L-
-
1-
D.
-2
^
^-^
-
u
k
f
a
5h
u
b
^^
r-
1
1
th r s
e n ch a
-4
.1 .2 .3 .4 .5 .6 .7 .8 .9 1jO 1.1 1.2 13 1/4 l.b 1.6 1.7 1A
TIME IN SECONDS
Fig. 6 — Melodic curves: "Joe took Father's shoe bench out"— sung.
-2.0
/\
-2.5
r-s
SPO^
'1
^EN-
,
\
\ â–
r
^
a.
/
r
"^V-
"V
u
J
o
\
•
•^ c
1>
^
\
i
*^
\
lUL
,
1/
— -t
^^
/''
1
^,
e —
^-
-J.U
6
J
\v
r.:
â– 'V
r-A-
r
SUNG
^!)
\
â– "â– ^
I
jv>
I-
a
\
A SPOKE
N
-3.£
\^
7 1J
.2 .3 .4 .5
TIME IN SECONDS
Fig. 7 — Melodic curves: "Joe took Father's shoe bench out"— spoken and sung.
356 BELL SYSTEM TECHNICAL JOURNAL
meaning of the words. Some of my friends in the musical profession
object to this statement of the situation but I think you will agree
that a singer's principal aim is to produce beautiful vowel quality and
to manipulate the melodic stream so as to produce emotional effects.
To do this, it is necessary in singing to lengthen the vowels and to
shorten and give less emphasis to the stop and fricative consonants.
It is for this reason that it is more difficult to understand song than
speech.
Characteristic Pitch or Frequency Levels for the Vowels
Now let us examine part of the speech wave of Fig. 1 in more detail.
Consider the vowel in the word "shoe."
The fundamental cycle was repeated 170 times per second. It is
evident that the second harmonic is very much magnified until it is
nearly as intense as the fundamental. In Fig. 8 is shown another
0.21 SEC.
Fig. 8 — Oscillogram of vowel u.
oscillogram of u intoned at 120 cycles per second. In this case the
3rd harmonic is magnified. An analysis of a number of u sounds
shows that components falling between 300 and 400 cycles per second
are always reinforced. This reinforcement is probably due to the
resonance characteristic of the mouth cavity.
Similar characteristic low pitch regions exist for the vowels in the
words, put, tone, talk, ton and father. A characteristic high pitch
region also exists for these sounds but the intensity of the components
falling in it are much less. For the vowels in the words tap, ten,
pert, tape, tip and team there are two characteristic regions of rein-
forcement which are of approximately the same intensity and which
are independent of the fundamental pitch. This is illustrated in
Fig. 9, which gives a spectrum analysis of the vowel "e" pronounced
at the four pitches indicated. The characteristic regions are at 375
cycles per second and 2400 cycles per second corresponding to pitches
— 1.4 octaves below and + 1.3 octaves above the reference pitch.
Experimental work ^ has indicated that for American speech the
characteristic pitch regions for the vowels and semi-vowels are those
shown in Fig. 10. For the first six vowels the components corre-
2 "Speech and Hearing," Harvey Fletcher, pp. 58, 59.
PHYSICAL CHARACTERISTICS OF SPEECH AND MUSIC 357
"E" AS IN "EAT"
PITCH - 128
1
1
1
11
"E" AS IN "EAT"
PITCH -170
1
1
1
1
1.
I
â–
ill
ll
â–
"E" AS IN "EAT"
PlTCH-192
â–
J
1
i
1
1 1
â–
.III
1
1
"E" AS IN *EAT"
PITCH-256
500 1000 1500 2000 2500 3000 3500 4
FREQUENCY
Fig. 9 — Spectra of "E" intoned at different pitches.
2
.^
i.
—
â–
â–
^
â–
^^m
^
â– ^
—
^^
^H
â–
^1
m^
.i
â– ^
—
^
1^
â–
B
â–
M
-1
^
^
-2
â–
^^
â–
â–
â–
-3
~5
3
a>
-ac
c
oj
a.
c
OJ
o.
£
>-
—
c
£
Fig. 10 — Characteristic resonance positions for the spoken vowels.
358 BELL SYSTEM TECHNICAL JOURNAL
sponding to the characteristic region of high pitch are much less
intense than those of low pitch. For the other vowels the Intensities
of both regions are about alike.
Oscillograms of the Unvoiced Continuants
Now let us examine more closely the wave forms for the fricative
sounds, s, sh, f, th. They are shown in Fig. 11. These show only
5H
^M
TH
Wir .V... 1 S. M . i- r ™ £i —
F
I I I I I I
.5 1
s
.5 1
Fig. 11 — Oscillograms of fricative consonants.
part of the oscillogram produced when each of these sounds was
continued for about one second. It is seen that these sounds contain
components having high pitches mostly above +1. It is seen that
they do not have the wave form repeated as uniformly as was the case
with the vowel sounds. They seem to be composed of a series of
explosions. For example, the oscillogram for "sh" looks very much
like one obtained from the sound of a sky rocket.
The f and th sounds are magnified six times in amplitude compared
to the sh and s sounds. Although much fainter they still show this
explosive character. There are 40, 45, 37 and 55 waves per each .01
second interval, respectively, for these four sounds corresponding to
4000, 4500, 3700, 5500 cycles per second.
PHYSICAL CHARACTERISTICS OF SPEECH AND MUSIC 359
Acoustical Power of Speech Waves
Keeping this picture before us, as to the physical composition of
speech, and its kinematic nature, let us now consider some statistical
averages. If ten different persons spoke the sentence discussed above,
there would be a considerable range of differences in the frequencies
and intensities used to transmit it through the air. To get a typical
cross-section of American speech, it would require at least 100 such
sentences pronounced by at least 5 men and 5 women. This would
involve the analysis of 18,000 fundamental sounds besides the transi-
tions between them. Also, as was seen from the oscillograms given
above, the wave form changes even where it is ideally supposed to
be constant so that three or four sample waves from each steady
state condition should be analyzed to find the components in each
sound. Thus, we have the problem of recording and analyzing about
70,000 such waves. To analyze such a wave by the usual academic
methods, namely, to plot the wave to a definite scale and then analyze
it into its components by means of a Henrici or similar analyzer, would
require at least two or three hours. So such a job for analyzing only
the steady-state part of speech would require about 210,000 hours, or
100 years working seven hours a day for 300 days per year. In other
words, such a method of attacking the problem is altogether too slow.
To find the average intensities and frequencies involved in con-
versational speech, much more powerful methods for obtaining
statistical averages were adopted.
There is a to and fro movement of the air particles simultaneously
with the alteration of the air pressure. When the source is so far
away that the disturbance can be considered as a plane wave, then
the following relations exist between the pressure p, the displacement
y, the velocity v, and the acceleration a of a layer of air particles, and
the frequency of vibration — , namely,
yco
voi = a, (2)
p = rv, (3)
where r is the radiation resistance of the air and is given by the product
of the air density by the velocity propagation of the wave. The
intensity / of the sound at any point is the power passing through a
square centimeter of the wave front and is given by
7=^- (4)
r
360
BELL SYSTEM TECHNICAL JOURNAL
If / is expressed in microwatts and p in bars, this reduces to
415
(5)
The intensity level / is defined by
/ = log.o / (6)
and is expressed in bels. These relations hold for any complex sound
as well as for a pure tone if p is interpreted as the root mean square
value of the pressure change.
It is seen then that all of these quantities can be determined by
making experimental measurements of the pressure change. For
accomplishing this the following methods were used.
/
FILTER
8,000
CYCLES TO
INFINITY
FILTER
8,000 TO
11,300
CYCLES
FILTER
5600 TO
8000
CYCLES
FILTER
4000 TO
5600
CYCLES
EIGHT
OTHER
FILTERS
FILTER
62.5 TO
125
CYCLES
FILTER
20 TO
62.5
CYCLES
I FLUX- \
I METER j
15-SECOND SYNCHRONOUS
CONTACTS MOTOR
FOR TAKING IS-SECOND
READINGS WHICH AVERAGE
ALL FREOUENCIES
Fig. 12 — Schematic of electrical circuit for measuring the average power-frequency
distribution of sounds.
The speech to be analyzed is picked up by a Wente condenser
microphone and sent into a vacuum tube circuit. This circuit is
arranged so that any one of 14 band pass filters can be inserted.
After passing through the filter the electrical speech wave is then
sent through a rectifier and finally into a meter. A schematic ^ of
^ See paper entitled "A New Analyzer of Speech and Music" by H. K. Dunn
(Bell Laboratories Record, November, 1930) and also paper entitled "Absolute
Amplitudes and Spectra of Certain Musical Instruments and Orchestras" by Sivian,
Dunn & White, Jour. Acotis. Soc. of America, Jan., 1931.
PHYSICAL CHARACTERISTICS OF SPEECH AND MUSIC 361
the circuit is shown in Fig. 12. Two kinds of meters are used. The
first is a flux meter as shown in Fig. 12 for integrating the speech
energy over any desired interval. When the rectifier is designed to
give a value which is proportional to the average voltage, then the
deflection of the needle of the flux meter will be proportional to the
average pressure times the time. In other words, this device will
read the average pressure during any desired time interval. In this
AMPLIFIER
MESSAGE
REGISTERS
SYNCHRONOUS
MOTOR
Fig. 13— Schematic of electrical circuit for measuring the peak power-frequency
distribution of sounds
way it is possible to find the average pressure in any one of the 14
bands. If the rectifier is adjusted so that the reading is proportional
to the square of the impressed voltage then the reading will correspond
to the average power. Knowing the calibration "* of the transmitter
* "Speech and Hearing," page 305, and also paper entitled "Absolute Calibration
of Condenser Transmitters" by L. J. Sivian, Bell System Tech. Jour., Jan., 1931.
362 BELL SYSTEM TECHNICAL JOURNAL
and also its distance from the mouth of the speaker, it is possible to
calculate approximately the average speech power.
The other type of meter shown in Fig. 13 consists of a series of
parallel circuits, each containing an argon filled three-electrode tube
connected in such a way that in adjacent circuits the tube breaks
down and allows the passage of current for voltage levels which are
6 db (decibels) apart. Ten such circuits then cover a range of 54 db.
Fig. 14 — Photograph of the level analyzer.
In each of these circuits a relay and counter are connected so that for
each tube discharge the counter operates. In this way the number of
times the tube breaks down is automatically registered. The speech
wave coming from the rectifier is sent into this meter where the peak
values are measured; that is, the number of times the pressure exceeds
a value fixed by each of these circuits will be registered automatically
by the corresponding counter. The apparatus is arranged so that
every other 8th second interval is measured, the intervening interval
PHYSICAL CHARACTERISTICS OF SPEECH AND MUSIC 363
being required for resetting the apparatus. In F'ig. 14 an observer is
shown reading the message registers after a test has been taken.
The breakdown tubes are seen at the left and the filters at the right
mounted on relay racks.
It is thus seen that with this apparatus 1000 observations may be
recorded on a four minute conversation, the final results being read
directly from the series of counters.
By the use of this and similar apparatus the following results have
been obtained. The average conversational speech power is 10 micro-
watts or 100 ergs per second. About 1/3 of the time no sound is
flowing due to the pauses and the stops to form consonants so that
the average conversational speech power is about 50 per cent higher
than this value if the silent intervals are excluded. Some of the
speakers will use a greater and some a lesser speech power than this
average. In Table I are shown the results with a large number of
TABLE I
Relative Speech Powers Used by Individuals in Conversation
Region of Average Speech
Power
Per Cent of Speakers .
1/16
1/8
1/4
1/2
1
2
4
below
to
to
to
to
to
to
to
1/16
1/8
1/4
1/2
1
2
4
8
7
9
14
18
22
17
4
above
8
Speakers. It will be seen that about 7 per cent of the speakers will
use in conversation average powers less than 1/16 the average while
about 4 per cent will use powers which are from 4 to 8 times as much
as the average. This value of 10 microwatts per second is of course
for average conversational intensity. When one shouts as loudly as
possible, this average speech power is raised about 100 fold and when
one whispers about as softly as possible and still produces intelligible
speech, it is reduced to about 1/10,000.
For describing in greater detail the powers involved in speech, we
will define the terms Mean Speech Power, Phonetic Speech Power and
Peak Speech Power. They are defined as follows:
The Mean Speech Power is the average speech power within any
one one-hundredth of a second period.
The Phonetic Speech Power is the maximum value of the mean
speech power of a fundamental vowel or consonant.
The Peak Speech Power is the maximum value of instantaneous
power over the interval considered.
364
BELL SYSTEM TECHNICAL JOURNAL
It was seen from the oscillographs that the vowels have much greater
phonetic powers than the consonants. Studies of these phonetic
powers for average conversation have indicated that for a typical
speaker they are as shown in Table II. The most powerful sound is
TABLE II
o'
680
u
310
ch
42
k
13
a
600
1
260
n
36
V
12
o
510
e
220
]
23
Ih
11
a'
490
r
210
zh
20
b
7
6
470
1
100
z
16
d
7
u
460
sh
80
s
16
P
6
a
370
ng
73
t
15
f
5
e
350
m
52
g
15
th
1
the vowel in the word "awl" which carries about 900 times as much
power as the weakest sound which is th as in thigh. This most
powerful vowel when intoned without emphasis is about 50 micro-
watts. The relative position in this table depends upon the emphasis
given. An emphasized syllable has about three times as much
syllabic power as an average one and as will be seen from the table
this is about the range of powers among the different vowels.
An analysis of a few oscillograms such as we first considered for
determining the peak powers was made and showed that the peak
powers are from 10-20 times the phonetic power. It is thus seen that
when the vowel in the word "awl" is emphasized, the peak power is
from 50 to 200 times the average speech power. To find how fre-
quently these peak powers occur, the apparatus described above using
the glow discharge tube circuits was used. The results obtained are
shown in Table III.
TABLE III
Per Cent of Number of db the Peak Power
1/8 Second in the Interval is Above
Intervals the Average Level
2 , . above 20
3 18 to 20
6 16 to 18
8 14 to 16
10 12 to 14
11 10 to 12
11 8 to 10
10 6 to 8
8 4 to 6
6 2 to 4
4 Oto 2
21 Below the average
PHYSICAL CHARACTERISTICS OF SPEECH AND MUSIC 365
These values confirm earlier results obtained by oscillographs and
give a much more detailed picture of the variation of the peak values
as the speech proceeds. About 2 per cent of the time the peak power
in l/8th second intervals exceeds the average power level by 20 db;
that is. it is more than 100 times greater. It is seen that a system
designed to transmit conversational speech of the best quality should
be capable of handling at least 1000 microwatts instead of 10 micro-
watts. It is also seen that the most frequently occurring peak is at
about 10 times the average speech power. For 21 per cent of the time
the peaks are below the average level. A large number of the l/8th
second intervals in this class are silent.
To find how the speech powers are distributed throughout the
pitch range similar measurements were made introducing successively
each one of the 14 band filters as indicated in Fig. 12. These bands
1/
-^
,^
^
N
::-:^
^,
ME
N/^
/
/'
"ST
-J
^
^
/
/
&.
\
\
/
/
lO
\n
/
WOMEN>
/
- o
s
\
/
/
\
\
/
_]
\\
/
-t
\
\
-2 -1
PITCH-P
OCTAVES FROM 1 KILOCYCLE
Fig. 15 — Distribution function for conversational speech.
SdP.
Pi
were arranged so as to cover about 1/2 octave pitch range except at
the two lower octaves where they cover a complete octave. From
the measurements on the average speech power in each band the curves
in Fig. 15 were constructed. They give the results for average con-
versational speech for both men's and women's voices. The ordinates
are such that the fraction of the total power F which is carried by any
pitch interval between Px and P2 is given by
Jp,
F = I 10^- dP.
Ipi
(7)
366 BELL SYSTEM TECHNICAL JOURNAL
In other words /3 is the intensity level per octave expressed in bels.
For example, the octave containing the most energy in men's voices
is — 1.75 to — .75 and it contains about lO"-^ or 31 per cent. The
octave below — 3 contains about 4 per cent and the octave above
+ 1 about 5 per cent. For women's voices these figures are 31 per
cent for the most intense region, which is the octave from — .85 to
+ .15, and .2 per cent and 7 per cent, respectively, for the other two
octaves.
Audible Pitch Limits
The audible pitch limits for conversational speech received at
various intensities are determined in the following way. It is seen
from Table III that the peak power exceeds the average power by
17 db 10 per cent of the time. The loudness of speech near the
threshold is probably determined by these louder components. For
convenience the term "effective intensity level" will be used when
speaking of these components only. With this nomenclature the
effective intensity level is 17 db above the average intensity level.
Using these figures and assuming that three-fourths of the speech
power is radiated through the hemisphere in front of the speaker,
then one can calculate that the effective intensity at one meter's
distance will be 6 X 10"^ microwatts per square centimeter or at an
effective intensity level of 22 db below one microwatt.
To determine the sensation level the pitches and intensities of the
components in the vowels must be considered. A study of the fre-
quency spectra of these vowels indicates that the loudest component
contains from 1/2 to 1/5 of the total power of the vowel. From this
it is concluded that the components determining the threshold are
from 3 to 7 db below the effective level of the speech. The threshold
of hearing for pure tones in the pitch region between — 1 and + 1
octaves is from — 85 to — 95 db with an average value of — 91 db.
Consequently, it is concluded that at the threshold the effective
intensity level for the speech is approximately — 86 db and the average
level approximately — 103 db. Since the effective level of the speech
at one meter's distance was shown to be — 22 db, it is seen that the
sensation level at one meter's distance is 64 db. If the speech wave is
uninterrupted by reflections then this level decreases 6 db when the
distance between the speaker and the listener is doubled. This level
will be raised or lowered in accordance with the intensity of the speak-
ing, the variation for different speakers being in accordance with the
data in Table I.
For example, using these relations one finds that the most probable
PHYSICAL CHARACTERISTICS OF SPEECH AND MUSIC 367
average speech power used by a person in conversation is 5 micro-
watts. The most probable sensation level of such speech at 1 meter's
distance is 61 db, at 10 meters' distance it would be only 41 db and
could be brought back to level of conversational speech at one meter's
distance only by the speaker shouting as loudly as possible.
If we use the peak voltmeter as shown in Fig. 13 and make measure-
ments upon the peaks in l/8th second intervals in each of the half
octave bands the results will be as represented by the curves of Fig. 16.
1
MAXIMUM
^
-^
■—
<.
y
^^
^
^
''i&%^
\
V
yy
X
\>
^^ .
i
^
^
<^
y
1
^\
\
\
X
^
^^
\
\\
/-
\
\
\
\'
\\
\
w
>
\V
\
\ —
V
A
Fig. 16 — -Peak levels for conversational speech (3 male voices), using 32 octave
average pitch intervals.
The top curves give the maximum level of the peak compared to the
average intensity. The other two give levels such that the peak
levels are below them 98 per cent, 90 per cent or 75 per cent of the time.
It will be seen that the most intense peaks occur in the pitch range of
— 1 to -f 1 octaves. In this pitch range the intensity levels of the
maximum peaks for the different components are approximately
the same, being 13 or 14 db above the average speech level.
It is interesting to note that in the higher pitch range the curves
in this figure are more widely separated than in the lower pitch range.
This illustrates an important characteristic of speech, namely, that
although components in the pitch range from zero to 2 octaves occur
which are just as intense as those in the lower range, they occur less
frequently. In other words, the spread in the intensities of the com-
368
BELL SYSTEM TECHNICAL JOURNAL
ponents which are successively occurring as the speech proceeds is
very much greater in the higher pitch regions.
As shown above, the threshold is determined for conversational
speech when the average speech level is at a — 103 db. For the same
reason that only 10 per cent of the peaks having the highest levels
determined the threshold for the speech as a whole, the curves labelled
90 per cent of this figure can be used as a basis for determining the
sensation level in each of the bands. When the ear of the listener
is 10 centimeters from the mouth of the speaker the sensation level
will be 84 db and the average intensity level will be — 19 db. If ao
9
^
/^
i :3
\
/
. 1
s
\
/
f _)
UJ
1 >
\
/
z
1 o
\
V
/
1 1-
1 J
\
/
/
1 '^
to
\
/
\
\
~'
\
~l
>
-1
PITCH
'
>
'
\
4
2 OCTAVES FROM I KILO-CYCLE 5
O O
o o
2 -i.
Fig. 17 — Speech audibility curve (male voices).
is the average threshold level for tones in each of the half octave
bands, then, if we subtract ao — 19 from each ordinate of the curve
in Fig, 16, we will obtain the sensation level of each half octave band.
A curve constructed in this way will be called an audibility curve and
is given in Fig. 17. This curve is for the case when the lips of an
average male speaker are 10 centimeters from the ear of an average
listener. It will be seen that the half octave bands above 3.25 octaves
and below — 4.25 octaves are just audible. If the distance between
speaker and listener is increased to one meter, which is the most
commonly used distance, then the audibility curve would be one which
is lowered 20 db from that one shown in Fig. 15 and the audible limits
would be + 3 and — 3.5 octaves, corresponding to frequencies of
PHYSICAL CHARACTERISTICS OF SPEECH AND MUSIC
369
8000 c.p.s. and 90 c.p.s. Similarly, if the distance is increased to
100 meters, the limits will be found to be + 1.85 and — 1.55 octaves.
These relations are true only when no other sounds are present.
Similar limits are easily determined when the listener is in the presence
of any other sound whose noise audiogram is known. In that case,
the ordinates in the audibility curve are reduced by an amount equal
to the corresponding ordinate in the noise audiogram.
These values are such that any half octave by itself within the
pitch limits will transmit audible sounds. This does not necessarily
imply that, when the undistorted speech is acting upon the ear, such
a half octave will transmit sounds whose presence can be detected.
To test this point several observers listened to speech reproduced by
a high quality loud speaker system which would reproduce all fre-
quencies from 40 to 15,000 uniformly and into which filters could be
introduced. These filters limited at desired cut-off positions the upper
and lower frequencies which were reproduced.
A large group of observers then listened to this reproduced speech
and they were asked to judge which was filtered and which was
unfiltered. The results of such tests are shown in Fig. 18. The
100
MALE
/
/
/^
1
Z
N
A
FEMALE
/
/
4
MALE
EECH
1 -^
eo°^
MAL
SPEE
-\
SPEE
CH
/
/
/ ^^
1 u CD
7n '-'°
-"\
\
/
/
1 °^ *-
UJ U
a bj
\\
^
/
/
50 -
^
I
PITCH
+2
Fig. 18 — Audible pitch limits for conversational speech.
ordinates give the per cent of correct observations and the abscissae
the cut-off frequency of the filter. Taking a 60 per cent correct
judgment as a criterion for determining the detectable pitch limits,
then it will be seen that the lower limit is — 3.5 octaves and the upper
limit 3.25 octaves for male speech which agrees with the results taken
from the audibility curve established directly from power measure-
ments upon speech and the threshold of hearing as described above.
For female speech the limits are — 2.9 and -\- 3.4 octaves. Sum-
marizing, then, it is seen that the most powerful components carrying
conversational speech, which are of any practical importance, are
about 4000 or 5000 microwatts while the principal components in
370 BELL SYSTEM TECHNICAL JOURNAL
the weakest sound carry only about l/20th of a microwatt. Even for
an extremely loud shout or for the most intense singing the maximum
power will not exceed more than about 100 times these values; that is,
they will not exceed 1 watt. The pitch range necessary for faithfully
transmitting men's and women's speech is from — 3.5 to + ?>.Z
octaves or from 90 to 10,000 cycles per second.
Acoustical Power Produced by Musical Instruments
Now we will look briefly at some of the same results obtained for
music by the use of some of these same measuring tools. In Fig. 19
^^S^^';Vw'V^^',^'s^ LOW E (dz) 147 CYCLES
)A'^^W\v<,'w^-V''XV','w^a.',\ low G (fa) 175 CYCLES
â– *Vw^'V^Vw^'V^*Vv,"^^^W''*^\v LOW CCbz) 233 CYCLES
Vv^;^S^;'>V^/^Vv\;\Vv^;'> THROAT E (da) 294 CYCLES
'.»At, -,Ai, <fi'u ','i''. .,<•. '
uyM^:;J>:J'^::J^^'yy;^^' throat g (fj) 349 cycles
'^>)>)>^'y'>)>}>}>^ MEDIUM C (b^) 466 CYCLES
^':j":X!J''J!''J!'j}'J^^^^^^ medium E (d^) 587 CYCLES
V'V^V'VVV,V)VV.V,V,V.V) MEDIUM G (f4) 698 CYCLES
it'
;ji M >t «\ n n,n >( i\ i{ ii »\ n M M M I
^ .' HIGH C Cb^) 932 CYCLES
Fig. 19 — ^Major triads of B-flat clarinet.
are shown typical waves produced by the clarinet. A complete
oscillogram of the waves produced when the instrument played its
full range of three octaves on the chromatic scale was taken. The
simple waves shown in the figure are those corresponding to the major
triad in each of these octaves. The entire record was about 250 feet
long. Such musical tones have a much more uniform wave form than
those from the voice.
PHYSICAL CHARACTERISTICS OF SPEECH AND MUSIC 371
The measurement of the peak power from typical musical instru-
ments used in an orchestra gave the following results.*
TABLE IV
Peak Power of Musical Instruments (Fortissimo Playing)
Instrument Peak Power in Watts
Heavy Orchestra 70
Large Bass Drum 25
Pipe Organ 13
Snare Drum 12
Cymbals 10
Trombone 6
Piano 0-4
Trumpet ^••'
Bass Saxophone 0-^
Bass Tuba 0-2
Bass Viol O-lo
Piccolo 0-08
Flute 006
Clarinet 0.05
French Horn 0.05
Triangle 0.05
The most powerful single instrument is the bass drum which gives
powers which exceed 25 watts in successive l/8th second intervals
about 6 per cent of the time it is being played. A 75-piece orchestra
20
10
y
â– ^
â– "^^^^
AXIM
UM^
x^
^\
/
r
s.
AVEF
AGE
K^USK
:al f
>OWE
N
^wr
1
lU
/
/
\j
^
5
o
0.
J
M
/
\most
PRC
BABL
-E
^ 10
<
UJ
0.
/
/
/
â–
N
o
/
/
\
X
>
bJ
y
/
\,
7^
N
V
30
•
f>
_
4
-
1
-
Z
-
1
1
J
3
<
PITCH -OCTAVES
Fig. 20— Maximum and most probable peak levels for a 75-piece orchestra.
« These results and those in Fig. 19 were taken from a paper by Sivian Dunn and
White entitled "Absolute Amplitudes and Spectra of Certam Musical Instruments
and Orchestras," Jour. Acous. Soc. of America, Jan., 1931.
372 BELL SYSTEM TECHNICAL JOURNAL
playing with full Aolume will produce peak acoustic powers as great
as 70 watts.
When such an orchestra played the four different selections, the
maximum peak powers varied from 8 to 66 watts, but the average
powers were .08, .07, .07 and .13 watts, respectively. Hence the
variation of the average power from selection to selection was much
less than that of the peak power. Both the peak powers and also
the average powers for the orchestra are about 10,000 times the
corresponding powers for conversational speech. In Fig. 20 the curves
show how the peak power was distributed among the different pitch
bands for this 75-piece orchestra. The curves give the average values
for the four selections. The zero line corresponds to a power of
approximately 1/lOth of a watt. The levels correspond to that which
was obtained in the half octave band acting alone. Although the
maximum peak was 70 watts for the unfiltered music when the heaviest
piece was being played, the most probable peak value in any half
octave band is less than 1/10 of a watt except for the octave between
- 2 and - 1 octaves, where it is slightly higher than this value.
The distance between the two curves increases as you go to either
side of this octave which is approximately that between middle "C"
and the "C" above it. This indicates that the components in this
region are more nearly alike in intensity and occur more frequently
than in the other regions. The top curve indicates that from the
standpoint of maximum peak values the half octaves from — 2| to
+ 1| octaves are all about equally important. As the pitch of a
component goes below 2\ octaves, its intensity decreases rapidly as
indicated in the figure. Very intense peaks occur occasionally with
frequencies as high as 10,000 or 12,000 cycles.
To find the lowest level used in orchestral music a violin player was
asked to play as softly as is ever customary while playing before the
public. Its average power was found to be about 4 microwatts. It is
thus seen that the peak power from a large orchestra is about
20,000,000 times the average power produced by soft violin playing.
Audible Pitch Limits for Musical Sounds
Measurement of the detectable pitch limits was determined in a
way similar to that described for conversational speech. The results ^
for typical musical instruments are shown in Fig. 21. For comparison
the results for speech and some common noises are also included.
It will be seen that the lower limit for music is determined by the bass
' A more comprehensive report of this work will soon be given in a paper by W.
B. Snow.
PHYSICAL CHARACTERISTICS OF SPEECH AND MUSIC 373
tuba, the bass viol, and the kettle drum, and its value is about 40 c.p.s.
The upper limit is determined by the snare drum, the violin, and
the cymbals, and is shown to be about 15,000 c.p.s. Summarizing,
then, for music the range of pitches covered by the components is
ACTUAL TONE RANGE
Miliiimnii ACCOMPANYING NOISE RANGE
-CUT-OFF PITCH AT WHICH 80* OF THE OBSERVERS
COULD DETECT THE FILTER
TYMPANI
BASS DRUM
SNARE DRUM
14"CYMBALS
BASS VIOL
CELLO
PIANO
VIOLIN
BASS TUBA
TROMBONE
FRENCH HORN
TRUMPET
BASS SAXOPHONE
BASSOON
BASS CLARINET
CLARINET
OBOE
SOPRANO SAXOPHONE
FLUTE
PICCOLO
MALE SPEECH
FEMALE SPEECH
FOOTSTEPS
HAND CLAPPING
KEY JINGLING
.,nn
â– mil
-
" 1
mill
iiiiiiii
'
niiii
1
jiiini
mill
**
llllloll
niiiii
*
-1
PITCH
+3 +4
Fig. 21 — Audible pitch range for speech, music and noise.
from — 4.7 to + 3.9 octaves, corresponding to the frequency range
from 40 to 15,000 cycles per second. The intensity ranges from about
70 watts to 4 microwatts, corresponding to an intensity level range of
73 db going from the average level of the softest violin playing to the
peaks in the heaviest playing of a full 75-piece orchestra.
The Statistical Energy-Frequency Spectrum of
Random Disturbances
By JOHN R. CARSON
A mathematical discussion of the statistical characteristics of Random
Disturbances in terms of their "energy-frequency spectra" with applica-
tions to such typical disturbances as telegraph signals and " static ".
IN a paper entitled "Selective Circuits and Static Interference"
{B. S. T. J., April, 1925) the writer discussed the "energy-
frequency spectrum" (hereinafter precisely defined) of irregular
random disturbances extending over a long interval of time. In
view of our lack of even statistical information regarding static or
atmospheric disturbances the specification of the energy-frequency
spectrum, denoted by R(co), was necessarily qualitative, and it was
merely postulated that
" R{co) is a continuous finite function of co which converges to zero
at infinity and is everywhere positive. It possesses no sharp maxima
or minima and its variation with respect to aj(co = Irf), where it
exists, is relatively slow."
In a paper entitled "The Theory of the Schroteffekt," ^ T. C. Fry
deals with a similar problem, namely, the energy or "noise" absorbed
in a vacuum tube from a stream of electrons with random time dis-
tribution. His method of attack is widely different from that of the
present paper. In a more recent paper on "The Analysis of Irregular
Motions with Applications to the Energy-Frequency Spectrum of
Static and of Telegraph Signals" (Phil. Mag., Jan., 1929), G. W.
Kenrick, by making certain hypotheses regarding the wave-form of
the elementary disturbances whose aggregate is supposed to represent
static interference, and by applying probability analysis, arrives at
explicit formulas for the "statistical" or "expected" value of R{co)
for a number of different cases.
I
In the present paper the statistical or "expected" energy-frequency
spectrum i?(co) of random disturbances is investigated by a method
which is believed to be somewhat more general and direct than that of
Kenrick.2 The results are applicable to the Schroteffekt, telegraph
^ Jour. Franklin Inst., Feb., 1925.
' Kenrick's analysis is based on a formula derived originally by N. Wiener instead
of proceeding directly from the Fourier integral.
374
STATISTICAL ENERGY-FREQUENCY SPECTRUM 375
signals and similar disturbances. The writer, however, concludes that
their application to "static" or "atmospheric" disturbances is of
questionable value owing partly to our lack of the necessary statistical
information regarding such disturbances and also to the fact that they
cannot be expected to have the "quasi-systematic" characteristics
necessary to the application of probability theory.
The energy-frequency spectrum of a disturbance, as the concept is
here employed, will now be defined. Let a disturbance <E>(0 exist in
the epoch ^ t — T and let
F(ico) = r(co) -f /.SXCO)
= r $(/)e^"V/. (1)
Then, as shown in my paper referred to above,
1 /»«! rtT
^Jo Jo
The energy-frequency spectrum is defined by the equation
G(co) = Lim4^|F(fco)|2, (2)
so that
r G{u:)d(^ = Lim^ r ^"-dt. (3)
Jo T^^ ^ Jo
It is on this last equation that the physical application of the concept
of the energy-frequency spectrum rests; namely, that it determines
the mean square value of $(/), as the epoch T is made indefinitely great.
Its principal application in electrotechnics depends upon the further
fact that, if $(/) represents an electromotive force applied to a net-
work of impedance Z{iw), the mean square current /- absorbed by the
network is given by ^
P = Lim i r Pdt = r .^^j^do^, (3a)
T^^^X Jo kMI
We now suppose that the function or disturbance $(/) is composed
of a number N of elementary disturbances ; thus
Ht) = i:a,n<l>,nO - Un), (4)
^A somewhat more involved formula gives the mean power absorbed. See my
paper referred to in the first paragraph.
376 BELL SYSTEM TECHNICAL JOURNAL
the mth elementary disturbance being supposed zero until / = /,„.
If we now write
4>„,{t)e^"'dt, (5)
it is easy to show by the methods employed in my previous paper
that
1
A'- 1 .V
+ 2 X^ Z aradniCmCn + S^n^r) COS a)(/„ — /,„) (6)
TO=1 n=m+l
N-\ N
+ 2 1] X! aynOiniCmSn — 5,„C„) sin Cj(/„ — /,„) .
m=\ n=m-\-\
This is more compactly expressible as
1
+ 2E' i {0,nanâ– frn{i0:)â– fn(-ic^)e"^'''^-''^'}ne.^V.riâ– (6a)
Now, obviously, if the amplitudes ai, • • • , a,v and the wave form of
the elementary functions 0i, • • • , 0,v are specified, G(w) is uniquely
defined and determined by the preceding formula. This, however, is
not the case in the problem under consideration, where at best the
functions are specified only statistically by probability considerations.
Under such circumstances, when the problem is correctly set and
sufficient statistical information is furnished for its solution, we
introduce the idea of the statistical energy-frequency spectrum i?(co)
defined as follows:
The statistical energy-frequency spectrum Rico) is equal to the weighted-
average of G(w) for all possible values of G(w), the weighting being in
accordance with the probability of the occurrence of each particular
possible value.
For example, the statistical value of a function f(xi, X2, •••, Xn),
where the variables Xi, • ■• , .r„ are defined only by probability con-
siderations, is, in accordance with the foregoing definition,
/1 00 nx /'CO
I dXipi(Xi) • 1 dX2p2{X2) ■■■I dXnpniXn)-f{Xi, X2, • • -, Xn),
*J — aa U — ji *J — jn
where pm{.Xm)dxm is the probability that x^, lies between .v,„ and
Xm ~r O'Xm'
STATISTICAL ENERGY-FREQUENCY SPECTRUM 377
To apply the foregoing concept and definition of the statistical
value of a function to the problem at hand it is necessary to suppose
that the typical impulse /m(*w) is a function of co and certain parameters
Xi, X2, • • ', Xn, and that these parameters are statistically specified by
probability considerations. Thus we suppose that pm0^m)d\m is the
probability that X„, lies between X,„ and X,„ + dK,. G(co) will then be
a function of co and Xi, X2, • • •, X„, the amplitudes ai, • • •, ay being
regarded as parameters, when defined by probability functions. We
then have, in accordance with the foregoing,
d\ipi{\) â– d\2p2{^0
x f" — 00
Xoo
d\npni\n)C{cO, Xj, Xo, •••, X..^ (7)
â– 00
X
(9)
II
To apply the foregoing to the simplest possible case let us suppose
that the elementary impulses are all identical; Qi = 02 = • • • o.v = 1.
and that their distribution in time is purely random. With these
assumptions it follows at once from (6) that
V . , . . , ^ V- t ^, . . ,„l — cos uT „ .Qs
R{co) ^- f{io^) -^ + 2 •- |/Oco)P :^ , r-^ CO. (8)
If f{iO) 9^ 0, this has a singularity at co = ; however
Lim -=. ^''dt = R{(^)do:
= V I 4>-dt -^ u-\ I 4>dt
Here v -^ N/T = mean frequency of occurrence of the elementary
impulses. This formula is in entire agreement with Fry's results for
the Schroteffekt (I.e.).
To consider a somewhat more involved problem, we shall suppose
that the durations of the individual impulses and their amplitudes are
distributed at random. We further denote the probability that the
duration of any impulse, selected at random, lies between X and
X + dX by p{\)d'K. Correspondingly, q{a)da denotes the probability
that its amplitude lies between a and a + da. The durations and the
amplitudes are then the statistically specified parameters.
W^e now postulate that $(/) is an alternating series of impulses of
378 BELL SYSTEM TECHNICAL JOURNAL
the same tvave form; i.e.
*(0 = Z (- l)'"am0m(^ - tm),
1
<i>,n{t) = <A(/), < / < X;„
= / > x,„,
/m = Xi + X2 + ' • • + X/n-l,
and we denote the mean frequency of occurrence, NIT, by v.
Substitution in the preceding formulas and straight-forward opera-
tions give
R{o,)=-\ a\{a)da- \f{io:, \)\^-p{\)d\
^ Jo Jo
— r rag(a)JaT- f'/O/co, X)^(X)e*"VX • f f{- iw,\)p{\)d\
"â– L Jo J Jo Jo
XLimi-^L L (-1)"-H p{\)e'-^d\\ . (10)
If we write
r p{\)e'-^d\ = p(ia>) = p, (11)
Jo
we have by straightforward procedure
Limi^E E (-1)"- ^(X).-VX = ^ , ., . , (12)
jvr-^oo iV OT=1 n = m+l L Jo J ^ ^ Pl^'^'J
whence
7?(co) =-^ Pa2g(a)Ja f \f(io:, \)\'-p(\)d\
-Mra.^a^.al-l'Mlff] , (.3)
TT L Jo J i 1 + P^^"^) J Real Part
/(iw, X) = r <l>{t)e^"'dt = c(co, X) + w(a), X),
Jo
£/(co) = rii'io:, X)/>(X)e^"VX. (14)
Jo
F(co) = rf{-ic^,\)p{\)d\.
Jo
where
STATISTICAL ENERGY-FREQUENCY SPECTRUM 379
If, on the other hand, we suppose that the impulses, instead of
systematically alternating in sign, are equally likely to be positive or
negative, the double summation term of (9) vanishes and
R(^) = i: f" o-qia)da • f" |/(/a;, \)\'p(\)d\. (15)
^ J-oo Jo
This follows from the fact that the amplitude a is equally likely to
be positive or negative. Consequently the integration with respect
to da must be extended from - oo to + =o and, since by hypothesis
g(_ a) = g(a), it follows that
f
aq(a)da = 0.
To apply the preceding formulas to actual calculations, it is necessary
to know the function /(^co, X) and in addition the probability functions
involved. These latter may be supposed known from statistical data
or calculable on theoretical assumptions. For example, if we assume
that the times of incidence of the elementary disturbances are dis-
tributed entirely at random, the application of well-known probability
theory gives ^(X) = ve'"^.
A third case is of interest. Here, instead of postulating that the
termination of one impulse coincides with the start of the next (i.e.
Wi = ^m + Xm), we suppose that the times of incidence are entirely
unrelated, and that the amplitudes are equally likely to be positive
or negative. For this case the formula for R(u)) is formally identical
with (15).
Ill
The foregoing analysis will now be applied to deriving what repre-
sents more or less accurately the statistical energy-frequency spectrum
of telegraph signals. To this end we shall suppose that the elementary
disturbance may have any one of three possible values (all equally
probable), characterized by durations Xi, X2, X3 and amplitudes
ai, az, as. The corresponding spectra of the elementary disturbances
are then determined by the equations,
Mio:) = f ' Me^'^'dt, (16)
Jo
Miu:) = f ' ct>(t)e''^'dt.
Jo
380 BELL SYSTEM TECHNICAL JOURNAL
The application of the precediniy analysis to this case gives
Sir
plus the real part of
X (aifi(- ^'co) + a2f2(- i^) + Os/.^C- ^'o;)) (17)
X Lim {. e' E [I (e'"'' + <?'"'= + ^'"'0 ]"-'"- ' •
It is to be understood that the real part of the second term is alone to
be retained.
If we write
1 1 / V _ 1 V 1 — r"^'"! \
j^l.l^lAe -re -re )j \ - x\ N N 1-x J
and
Lim |. L E = 7-^: •^- <
A— »•« ^* i A.
= T * = 'â–
There is therefore an infinity at co = 0, as we should expect. Its
measure, however, is finite.
The preceding is merely an example which admits of extension to
more complicated types of signals, as will be obvious to the reader.
For example, the probabilities of the elementary signals need not be
the same and their number need not be restricted to three.
IV
In all the cases discussed above it will be observed that the dis-
turbance is "quasi-systematic" in the sense that the elementary
disturbances are all of the same wave-form differing only in duration
and amplitude. Indeed, some such assumptions as these are essential
to the application of the mathematical theory. In the case of atmos-
pheric disturbances we have no reason to suppose any such quasi-
systematic character exists. Furthermore, even if for the sake of
argument, we suppose that the elementary disturbances, which make
up static, have a common wave form at the point at which they
STATISTICAL ENERGY-FREQUENCY SPECTRUM 381
originate, they would vary widely in this respect after arriving at a
common receiver. The writer is therefore of the opinion that the
quotation from his previous paper appearing at the start of this article,
represents all that can safely be said regarding the spectrum of static
and that our present knowledge is insufficient to justify the application
of probability analysis to the problem. All that we can say is that
the part of R{o)) which contributes to "static interference" is simply
V \ ^
Lim -'-^jY. am^|/m0'w)|2,
jV-^^ tt iV 1
a result deducible from (6) and in agreement with the conclusion of
my original paper (I.e.) . It is here supposed that the times of incidence
are distributed at random. This formula, however, supplies no useful
information in the absence of data regarding the wave forms and
amplitudes of the individual disturbances.
Bridge Methods for Locating Resistance
Faults on Cable Wires
By T. C. HENNEBERGER and P. G. EDWARDS
In this paper are discussed bridge methods for locating resistance faults
on cable wires, with special reference to the theory of methods for (1) locat-
ing insulation faults which cause complete cable failure, (2) locating insula-
tion faults of high resistance, and (3) locating series resistance unbalances.
The methods described are better adapted to the toll than to the exchange
telephone cable plant, since they require that the conductor resistances of
the wires used for measurements be equal and, in general, that measurements
be made from each end of the faulty cable.
IN the toll telephone plant, insulation faults such as "grounds" and
"crosses" are usually located by the "Varley loop" method, which
involves essentially the measurement of the d.-c. resistance of the
faulty wire between the point of fault and one end of the cable, and
the comparison of this resistance with the total conductor resistance
of the wire to obtain the "percentage location" of the fault on a re-
sistance basis. Corrections are then applied to account for such
factors as the resistance of the leads between the cable and the bridge,
the resistances of loading coils, and non-uniformity of conductor
resistance caused by temperature differences between underground and
aerial sections of the cable. After all corrections are applied the
corrected percentage location is converted into distance from one
cable end to the fault.
In general, the most troublesome insulation fault to locate is a
"wet spot" due to absorption of moisture by the insulation through a
defect in the lead covering of the cable, which results in low insulation
resistance between wires and to ground. Standard apparatus now
available for locating grounds and crosses is sufficiently sensitive to
permit accurate locations of wet spots up to about five megohms in
resistance. The Varley loop methods ordinarily employed in con-
nection with the apparatus will give accurate results provided a wire
of very much higher insulation resistance than the faulty wire is used
as the "good" wire for measurements. These are the conditions
which usually are found when wet spots occur. Cases occur occa-
sionally, however, in which a "good" wire having sufficiently high
insulation resistance as compared to the faulty wire cannot be obtained,
either because all of the wires available for measurements are affected
382
BRIDGE METHODS FOR LOCATING RESISTANCE FAULTS 383
by the fault or because the fault resistance is high. The methods for
locating insulation faults discussed in this paper are especially applic-
able to such cases.
Resistance unbalances on cable wires are of relatively infrequent
occurrence and are usually difficult to locate. A method frequently
employed for locating such faults is to measure the impedance un-
balance at various frequencies of a circuit containing the faulty wire
and to analyze periodic impedance-frequency curves plotted from the
measurements. 1 The methods for locating series resistance unbalances
discussed in this paper involve the use of ordinary Wheatstone
bridges, are simple to apply, and give results which are believed to be
comparable to those obtained by the impedance-frequency method.
NoR:iiAL Insulation Resistance of Cable Wires
The values of insulation resistance obtained by measurements on
cable wires which are not faulty are dependent on the circumstances
in which the measurements are made. In the case of paper-insulated
telephone cable the most important factors affecting insulation re-
sistance are electrification period and temperature.
The following discussion of normal insulation resistance refers
particularly to measurements between wires of pairs in a typical
repeater section of aerial toll cable approximately 50 miles long, the
wires being at ground potential at the time of application of the
testing potential. Insulation resistance to ground is also of interest,
but is difficult to measure accurately in long lengths of cable because
of interfering potentials. As a rough approximation, normal insulation
resistance between a wire and ground can be considered to be about
two thirds as great as normal insulation resistance between wires.
A curve illustrating the variation of insulation resistance between
wires of a typical cable pair over a 30-minute electrification period is
shown in Fig. 1. In general, the electrification periods necessary for
obtaining reasonably constant values of insulation resistance differ
appreciably for different pairs, and for the same pair at different times.
The usual period ranges from 15 minutes to an hour for a pair 50 miles
long. Routine measurements are generally made, however, using
electrification periods of one minute.
The paper used for insulating the wires of telephone cable has an
appreciable negative temperature coefficient of insulation resistance.
This is indicated by the curve of Fig. 2 which shows variations of
average insulation resistance with temperature. The points for the
1 "Telephone Circuit Unbalances," by L. P. Ferris and R. G. McCurdy, A. I. E. E.
Transactions, 1924, Volume XLIII, page 133L
384
BELL SYSTEM TECHNICAL JOURNAL
curve were obtained by averaging, for each five-degree range of tem-
perature, the insulation resistances obtained by measurements made
??600
^/f
y^
f^
/
/
/
f
/
/
f —
Ele
1
ctrificatior
J 1
1 Per
J .
1 1
od -Minutes
J 1 1
Fig. 1 — Variation of insulation resistance with electrification period — -typical 50 mile
aerial cable pair.
E400
10 20 30 40 bO 60 70 80 90
Outside Temperature at Measuring Station -Degrees Fatirentieit
Fig. 2 — Variation of average insulation resistance with temperature — typical repeater
section of aerial cable.
daily over the course of a year on representative pairs in a repeater
section, using electrification periods one minute long. It has been found
that the percentage change in insulation resistance per degree change in
BRIDGE METHODS FOR LOCATING RESISTANCE FAULTS 385
temperature differs widely for different cable sections and even for the
individual pairs in a particular section. The average change per
degree Fahrenheit is probably about four per cent, for the temperature
range encountered in the plant.
15100
5200
250
225
; 200
.—
VO^
K
^J
y
\
S,
/
/
Vl
^\
r^
S ji
V
\
s,^
y
f
J
a/
<
>....
I
V
r
h^
y
r
\^
V
t*^"
\
S,
/
\
/
\-
r^
30
35
40'
45 Si
— 50 Q
Fig. 3— Variation of insulation resistance, loop resistance and temperature over
24 hour period — -typical 50 mile aerial cable pair.
The curves of Fig. 3 illustrate comparative variations of insulation
resistance between wires of a representative cable pair, conductor
resistance of the pair, and outside temperature, during a 24-hour
period which included a sunny summer day. The curves were plotted
from measurements made every half hour, one-minute electrification
periods being used when measuring insulation resistance. It is not
uncommon to find that the insulation resistances of particular pairs
vary by factors of three to one during the course of a day.
Comparative variations of average insulation resistance between
wires of pairs and of mean outside temperature over the course of a
year are illustrated by the curves of Fig. 4. The points for the insu-
lation resistance curve were obtained by measuring the insulation
resistances of a number of pairs each working day during the year,
using one-minute electrification periods, and averaging the measured
values for each day.
In general, average insulation resistance is likely to vary by a factor
of 15 to one during the course of a year. Individual pairs are, of
course, subject to much wider seasonal variations in insulation re-
sistance. During winter it is not uncommon to find particular pairs
in a 50-mile repeater section with insulation resistances between wires
386
BELL SYSTEM TECHNICAL JOURNAL
DEGREES FAHRENHEIT
MECOHMS
BRIDGE MRTIIOns FOR LOCATING RESISTANCE FAULTS 387
of several thousand megohms, while during summer, especially in
cables which have been in service for a number of years, the insulation
resistances between wires of some pairs in a 50-mile repeater section
may be as low as 25 megohms (1250 megohm-miles).
Varley Loop Method
The Varley loop circuits which are used ordinarily for locating
grounds and crosses on wires of toll cable are illustrated in Figs. 5 and
6. The Wheatstone bridge has equal ratio arms, A. The "good"
jVarley
Fig. 5 — Varley loop for grounds.
and faulty cable wires have equal conductor resistances, r, and are
connected together at the distant end of the cable. F is the resistance
of the fault, and x is the conductor resistance of the faulty wire between
the fault and the distant end of the cable.
iVarley faulty wire
Fig. 6 — ^ Varley loop for crosses.
388 BELL SYSTEM TECHNICAL JOURNAL
With the battery switch in "Varley" position, a Varley measure-
ment is made by balancing the bridge to a rheostat value, V, at which
there is no galvanometer current. Then:
A r -\r X
A r - X + V
x = |- (1)
It will be noted that the fault resistance, F, is in series with the
battery and has no effect on the measurement except to limit the
sensitivity of the bridge.
With the battery switch in "loop" position, a loop resistance
measurement is made by balancing the bridge to a rheostat value, L.
Then:
L
2
From these Varley and loop measurements the percentage location
of the fault, on a resistance basis, can be calculated as follows:
V
From the distant end: — (100 per cent).
L — V
From the measuring end: — j (100 per cent).
Corrections for resistances of bridge leads, loading coils, etc., are then
made, the corrected percentage location is converted into feet, and the
location of the fault is determined by reference to cable records.
These Varley circuits and formulas are well adapted to the toll
cable plant where wires are usually well balanced in conductor re-
sistance, and the resistance of the leads between the bridge and the
cable is small compared to the conductor resistance of the cable wires.
In exchange cable work, modified forms of the Varley loop, which do
not require that the "good" and faulty wires be of equal conductor
resistance and which correct automatically for the resistance of bridge
leads, are frequently used.
Total Cable Failures
In the case of total cable failure, due, for instance, to a wet spot,
there are no wires in the cable which are unaffected by the fault, and
the fault resistances of a large number of the wires are low, i.e., of the
same order of magnitude as the conductor resistances of the wires.
BRIDGE METHODS FOR LOCATING RESISTANCE FAULTS 389
Two methods by which such faults can be located are discussed below:
A "Corrected Varley" method which may be used provided two wires
having fault resistances to ground differing by at least 25 per cent are
available for measurements; and a "Straight Resistance" method
which does not require that the two wires have faults of unequal
resistances.
Corrected Varley Method
Consider a cable in which all wires have low insulation resistance to
ground because of a wet spot, and assume that from among the faulty
wires two wires are selected for a Varley measurement. Assuming a
bridge having equal ratio arms, A, the Varley network can be repre-
sented as shown in Fig. 7, where M and F are the effective resistances
of the faults on the two wires, r is the conductor resistance of either
wire, and x is the resistance of that portion of either wire which is
between the distant end of the cable and the faults.^
I
Fig. 7 — Schematic circuit — corrected Vailey method.
The Varley circuit of Fig. 7 is equivalent to the Varley circuit of
Fig. 8, where the "tt" type network formed by the three resistances,
M, F and 2x, has been replaced by a "T" type network having resist-
ance values as indicated. When the bridge is balanced by adjustment
of the rheostat to a resistance, V, at which there is no galvanometer
2 The actual faults form a "tt" type network consisting of a ^resistance between
wires and a resistance between each wire and ground. The "tt" type network has
been replaced by a "T" type network having resistances, M and F, between the two
wires and the branch point of the network, and a third resistance connectmg the
branch point to ground. This third resistance is in series with the bridge battery
and its only effect is to limit the sensitivity of the bridge. To simplify discussion
the resistances, M and F, are shown connected directly to ground, and the third
resistance is considered to form a part of the resistance shown connected between the
battery and the junction point of the ratio arms of the bridge.
390
BELL SYSTEM TECHNICAL JOURNAL
current :
x +
2Mx
M -\- F+ 2x
r — X -\-
2Fx
M + F -\- 2x
+ V.
Solving for x:
V {M + F)
2 (M
V)
(2)
Comparison of this formula with Formula (1) indicates that the
factor V/2, as determined by Varley measurement, represents the
r-x
r-x
"1 M+F + 2x ^
I
Fig. S^Equivalent circuit — corrected Varley method.
apparent rather than the true resistance between the distant end of
either wire and the location of the faults. The factor . . ,"_ p _ y\
is a correction factor and expresses the relation between V/2 and
the true resistance, x. If the fault, M, is very much higher in resist-
ance than either the fault, F, or the balancing resistance, V, the
correction factor becomes practically equal to one and V/2 becomes
practically equal to x. In these circumstances the wire having the
fault, M, can properly be called a "good" wire and Formula (1) will
give accurate results.
Since the apparent resistance, V/2, can be determined by \'arley
measurement the faults can be located if the value of the correction
factor can be determined. The correction factor can be evaluated
by additional measurements made on the two faulty wires from the
end of the cable opposite to that used for the \'arley measurement, as
described below.
Referring to Fig. 7, the resistance of either wire between the faults
and the end of the cable opposite to that used in making the Varley
measurement is x. If a loop resistance measurement is made from
BRIDGE METHODS FOR LOCATING RESISTANCE FAULTS 391
this opposite end, using a bridge having equal ratio arms, 7i'itli the
distant ends of the wires open, and the resistance in the bridge rheostat
at balance is designated Lq:
il/ + F = Lo - 2.r.
If a Varley measurement is made from the same end, using a bridge
having equal ratio arms, it'ith the distant ends of the ivires open, and the
resistance in the bridge rheostat at balance is designated Fq:
M - F = Fo.
Substituting these values of {M + F) and {M - F) in (2) :
x=^^. (3)
"" 2 Fo ^ ^
Application: To apply the Corrected X'arley method, an ordinary
Varley measurement is made from one end of the cable, and additional
loop resistance and Varley measurements, as described above, are
made from the opposite end. The values of balancing resistance thus
obtained are substituted in Formula (3). The location of the trouble
on a resistance basis, x/r, can then be calculated, and the location can
be converted into feet in the usual manner.
Usually it will be necessary to determine the loop conductor re-
sistance, 2r, of the faulty wires from cable records rather than by
measurement at the time the location is being made. A measurement
of loop conductor resistance would be in error because of the low
resistance shunt {M -f F) on the portion of the loop between the faults
and the short-circuited ends of the wires. The accuracy of location
is dependent, therefore, on the accuracy to which conductor resistance
can be estimated.
In cases where it is desirable to use the Corrected Varley method
the fault resistances will be low, so that usually the balancing resist-
ance, Lo, will not exceed 10,000 ohms. If Lo is too high to measure
using a bridge with equal ratio arms, unequal ratio arms, A and B,
A
may be used and the quantity -^ Lo substituted for Lq in Formula
(3). Measurement of Fo, however, should be made using a bridge
with equal ratio arms.
The Corrected Varley method will give accurate results only under
the following conditions:
(1) Both faults must be at the same point along the cable.
(2) The fault resistances must remain constant throughout the test.
392
BELL SYSTEM TECHNICAL JOURNAL
(3) The resistance of the fault on one wire must be higher than the
resistance of the fault on the other wire.
(4) The conductor resistances of the faulty wires must be equal.
In the practical application of the method, care must be exercised
in selecting the wires to be used for measurements. The resistance,
M, of the fault on the wire which is connected to the ratio arm of the
bridge when measuring V should be appreciably higher (at least 25
per cent higher) than the resistance, F, of the fault on the wire con-
nected to the rheostat arm of the bridge. This can be understood by
considering that as M and F approach each other in value the correc-
tion factor becomes larger and the Varley balancing resistance, V,
approaches zero, i.e., the apparent location of the trouble approaches
the distant end of the cable. Errors in measurement become in-
creasingly important as V and Fo become smaller.
Accurate results will not be secured if the resistances of the faults
vary while a set of measurements to determine V and the correction
factor is being made. It is advisable, therefore, to make a number of
separate sets of measurements, and to base the location on those sets
which appear to be consistent.
Straight Resistance Method
In many cases of complete cable failure the faults on all of the wires
are of practically equal resistance, and the Corrected Varley method
cannot be used successfully. The Straight Resistance method de-
scribed below has the advantage that the wires used for measurement
need not be unequal in fault resistance.
Schematic circuit — straight resistance method.
The Straight Resistance method is based on the assumptions that
the wires on which the tests are made are of equal conductor re-
BRIDGE METHODS FOR LOCATING RESISTANCE FAULTS 393
sistance, that the fault resistances are comparable in magnitude to
the conductor resistances, and that the fault resistances remain
constant while a set of measurements is being made.
Referring to Fig. 9, assume that, from among the faulty wires, two
wires are selected having a fault of low effective resistance, {M + F),
between wires. Let r be the conductor resistance of either wire be-
tween cable Ends 1 and 2 ; and let {r - x) and x be the conductor
resistances of either wire from Ends 1 and 2, respectively, to the fault.
With the wires open at End 2, the resistance between wires is
measured at End 1 by means of a bridge having equal ratio arms and
arranged for an ordinary loop resistance measurement. Calling the
rheostat resistance at balance, Loi:
Loi = 2{r-x) + 0/+F).
Similarly, with the wires open at End 1, the resistance between wires
is measured at End 2. Calling the rheostat resistance at balance, L02:
L02 = 2x + (.1/ + F).
Combining the equations for Loi and L02:
Z/02 — Loi ~ 4.A; — If.
and therefore:
^ = 2a- + (Los - Loi) ^ ^^^
4
N 2/- - (L02 - Loi) (z^
{r - x) = ^ • l^J
Application: The Straight Resistance method involves only simple
resistance measurements, Loi and L02, from the two ends of the cable.
The loop conductor resistance of the faulty wires is obtained from
cable records. The values thus secured are substituted in Formula
(4) or (5), and the location, x or {r - x), is converted into feet in the
usual manner.
Since the conductor resistances of the faulty wires must be equal,
measurements should be made on the two wires comprising a pair
when practicable. The effective fault resistance between wires should
be low; otherwise slight errors in measurement might cause large
errors in calculated location. However, in cases where the fault re-
sistances are too high to be measured using bridges with equal ratio
arms, unequal arms, A and B, may be used and the quantity
^ (L02 - Loi) substituted for (L02 - Loi) in the formulas.
X5
394 BRLJ. SYSTEM TECHNICAL JOURNAL
In connection with botli the Corrected \'arley method and the
Straight Resistance method, it is possible to modify the measuring
schemes and obtain somewhat more compHcated formulas for the
location of the faults. The specific measuring schemes which have
been described are those which it is felt are most practicable for fault
locating work on toll cable.
Insulation Faults of High Resistance
In order to locate faults of high resistance, sensitive galvanometers
and highly insulated bridges must be employed, and the fault locating
methods must correct for factors peculiar to the locating of such faults.
If the resistance of the fault is high enough to be comparable in mag-
nitude to the normal insulation resistance of the faulty wire, the effect
of normal insulation resistance must be taken into account. In the
case of a high resistance wet spot, it may happen that all wires in the
cable are affected to some extent by the fault so that no wire of high
insulation resistance compared to the selected faulty wire is available
for measurements.
The solutions of the \^arley networks for high resistance faults are
more readily obtained by approximate than by exact mathematical
reasoning, and will be worked out by the process of combining all of
the "effective faults" on the wires into a single resultant fault and then
solving the bridge network for this fault. The approximate solution
is based on a principle which for the purposes of the present discussion
can be stated as follows:
Any two shunt faults of high resistance along a ware can be replaced
by a single resultant shunt resistance located between the two
faults at a point the distance of which from either fault is
directly proportional to the fault resistances.
Thus, if M and F are the resistances of two faults at separated points
along a wire, and m and / are their respective distances from the re-
sultant, then:
M _ m
~F~7'
The application of this "Rule of Resultant Faults" to Varley
measurements can be shown as follows : Let M and F be the effective
resistances of the faults on two cable wires at the same point along the
cable; let r be the conductor resistance of either wire between the
cable ends, and x the resistance of that portion of either wire which
is between End 2 of the cable and the faults. Let V be the value of
balancing resistance for a Varley measurement made from End 1,
using a bridge with ecjual ratio arms, as indicated in Fig. 10.
BRIDGE METHODS FOR LOCATING RESISTANCE FAULTS 395
Applying the Rule of Resultant Faults, the apparent location of the
faults as determined by the Varley meeisurement will be at a point
between the two faults, and at a distance from either fault which is
directly proportional to the fault resistances. Let c be the resistance
End 2
I
Fig. 10 — -Location of a resultant fault.
of the portion of the wire between the distant end of the cable and
the apparent location. Then:
M
F
X + c
X — c^
M - F
C — X
M + F
When the bridge is balanced for the Varley measurement:
V
' = -2'
Equating the two values of c and solving for x:
_ V M + F
^ ~ 2 M - f'
(6)
Comparison of Formula (6) with the more exact Formula (2) for
the same case indicates that the Rule of Resultant Faults will give
accurate results only if the fault resistances are high compared to the
conductor resistances, and if M is of appreciably higher resistance than
F.
If M equals F, the location will be indeterminate: The two faults
will have no effect on the balance point of the bridge and V will be
zero.
396
BELL SYSTEM TECHNICAL JOURNAL
Double Varley Method ^
The distributed normal insulation resistances of cable wires can be
considered, in so far as fault locating measurements are concerned, as
though they were single resistances concentrated at some point along
the wires (Rule of Resultant Faults). Consider two wires having
equal and correspondingly distributed normal insulation resistances,
N, which appear to be concentrated at some point b ohms from End 2
of the wires, and assume faults of effective resistances, M and F, on
the wires at a point x ohms from End 2. Let r be the conductor
resistance of either wire, and Vi and F2 the balancing resistances for
Varley measurements from Ends 1 and 2 of the wires, respectively,
using bridges with equal ratio arms as indicated in Fig. 11.
Fig. 11 — -Schematic circuit — -double Varley method.
Applying the Rule of Resultant Faults, let Ci be the apparent loca-
tion, in ohms from End 2, of the resultant of M and TV, and let C2
be the corresponding location of the resultant of F and A''. Then:
M
N
and correspondingly:
Cl
c<>
Ci — X
b — Ci'
Alb + Nx
M -\- N
Fb-\- Nx
F+ N
The equivalent resistance of the resultant of .1/ and N is MN/M + N.
and of the resultant of F and N is FN/ F -\- N. Let cs be the apparent
'The Double Varley method has been described in "Cable Testing," a paper
read by E. S. Ritter before the Nottingham Centre of the Institute of Post Office
Electrical Engineers (British), May 25, 1922. In that paper it is stated that the
method is due to Mr. H. T. Werren.
BRIDGE METHODS FOR LOCATING RESISTANCE FAULTS 397
location, in ohms from End 2, of the resultant of these two resultants,
as indicated in Fig. 12.
Fig. 12 — ^Equivalent circuit^double Varley method.
Again applying the Rule of Resultant Faults:
_ Nx{M - F)
^' ~ M{F + iV) + F{M + N) '
For the Varley measurement from End 1 of the cable:
cz =
V,
Equating these two values of Cs and solving for x:
X =
Fi
il/+ F
+
2MF
M - F ' N{M - F)
Likewise, for the Varley measurement from End 2 of the cable:
M + F , 2MF
X = r
2
M - F ' N{M - F)
(7)
(8)
By equating the two values of x found in (7) and (8), the value of the
"correction factor" for the Varley measurements can be determined:
M + F
IMF
2r
M - F ' N{M - F) Fi + F2
Substituting this value of the correction factor in Formula (7)
X =
Fi+ F2
(9)
398 BULL SYSTEM TECHNICAL JOURNAL
Likewise, the resistance of one wire between l^nd 1 of the cal)Ie and
the faults is:
i' - '') = v^,- ('")
AppUcalion: To apply the Double Varley method, ordinary \'arley
measurements, V\ and V-i, are made from the two ends of the cable,
using bridges with ecjual ratio arms, and the loop resistance, 2r, of
the wires is measured. The location, x or {r — x), can be calculated
from Formula (9) or (10), and then converted into feet in the usual
manner.
Similarly, using the Rule of Resultant Faults, it can be shown that
Formulas (9) and (10) also apply when only one of the wires used for
Varley measurements is faulty. In this case the resistance, x, of the
portion of the faulty wire between the distant end of the cable and
the fault is:
V F
where V is the balancing resistance for a Varley measurement made
from one end of the cable. This formula indicates that, w^here the
ordinary Varley method (Figs. 5 and 6) is used, the insulation re-
sistance of the "good" wire should be at least several hundred times
as high as the fault resistance of the faulty wire. If this condition
does not obtain the Double Varley method should be used. It will
be clear, however, that the Double Varley method may be used, if
desired, instead of the ordinary Varley method in cases where a wire
of suflficiently high insulation resistance to be a "good" wire is avail-
able. In such cases the sum of the Varley balancing resistances ob-
tained by measurements from the two ends of the cable will be equal
to the loop resistance and Formula (9) will reduce to Formula (1).
The Double Varley method is workable only if the conductor re-
sistances of the two wires used for measurements are equal. It can
be shown that, if the conductor resistance of the ware having the fault,
AI, is rm and that of the wire having the fault, 7^, is r/, and if the normal
insulation resistances of the wires are equal and uniformly distributed
so that they may be regarded as concentrated at the middle of each
wire. Formula (9) becomes:
X = r/
^^ 12MF + N{M -f F)-\ -f '-^-^ IMF + N{M -f F)]
Yl±Il[_2MF + N{M ^ F)-]
+ {rj - rm)lMF + N{M + F)] J
BRIDGE METHODS FOR LOCATING RESISTANCE FAULTS 399
As indicated by the above discussion, the hniitations of the Double
Varley method are as follows:
1. There must be only one actual fault on any one cable wire.
2. The fault resistances must remain constant throughout a set of
measurements to determine Fi and V^-
3. If b(-th of the wires used for the Varley measurements are faulty,
tlie faults must be at the same point on each wire, the re-
sistances of the faults must be unequal, and the resistance
of the fault on at least one of the wires must be high compared
to the conductor resistance of the wire.
4. If the fault resistances are high enough to be comparable in
magnitude to the normal insulation resistances of the faulty
wires, the normal insulation resistances must be equal, and
correspondingly distributed along the wires.
5. The conductor resistances of the wires must be equal.
It will be understood that since the Double Varley method is ap-
plicable only when the resistance of the fault, M, is high compared to
the conductor resistances of the wires, the Corrected Varley method
or the Straight Resistance method should be used in cases where M
is comparable in magnitude to the conductor resistances.
Series Resistance Unbalances
The methods for locating series resistance unbalances discussed in
this paper involve essentially the balancing of the faulty wire against
a "good" wire of equal capacitance by adding resistance to the "good"
wire at the testing end until the effective impedances of the two wires
are equal. A simple relationship then exists between the balancing
resistance required, the resistance of the fault, the length of the faulty
wire between the distant end of the cable and the fault, and the total
length of the faulty wire. The circuit arrangement used depends on
whether the cable under test is long or short.
The circuit arrangement for applying the test to short cables is
shown in Fig. 13.
T
Audible
frequency
Shielded-ir
Transformer
Fig. 13 — Schematic circuit — short cable method for locating a series resistance
unbalance.
400
BELL SYSTEM TECHNICAL JOURNAL
The wires 1-2 and 3-4 form the pairs of a quad containing a series
unbalance of resistance, F. The total length of the faulty wire is T,
and the length of the portion of the faulty wire between the distant
end of the cable and the fault is D. The bridge has equal ratio arms,
A, and a balancing resistance, R. The audible frequency generator
is a buzzer or other source of relatively low frequency current.
The bridge is balanced first with the distant ends of wires 1, 2, 3
and 4 open, and then with the distant ends of wires 1, 2, 3 and 4 con-
nected together. The location of the unbalance from the distant end
can be calculated from the formula:
D = T
Re
where Ro and Re are the balancing resistances for the measurements
with the distant end open and the distant end short-circuited, re-
spectively. This test is suitable for use only on non-loaded cable, up
to a few miles in length.
Reversing
switch
Fig. 14 — Schematic circuit — long cable method for locating a series resistance
unbalance.
The bridge arrangement for applying the test to long (either loaded
or non-loaded) cables differs from that for short cables in that the wires
of each pair, 1-2 and 3-4, are connected together at the distant end
when measuring Rq, and a testing current of very low frequency is
used. A battery, reversed either manually or by means of a motor-
driven commutator, provides a satisfactory source of current, as in-
dicated in Fig. 14.
With the wires of each pair, 1-2 and 3-4, connected together at the
distant end as shown, the balancing resistance is adjusted to a value
BRIDGE METHODS FOR LOCATING RESISTANCE FAULTS 401
Rq at which no deflection of the galvanometer occurs when the battery
is reversed. The two short-circuited pairs are then connected to-
gether at the distant end, the reversing switch is left in one position,
and the rheostat is adjusted to a value Re to balance the bridge. The
location of the unbalance from the distant end is:
D = T
Ro
Re
As will be clear from the following discussion, both the formula for
the short cable method and that for the long cable method are based
on the assumption that the wires under test are of short electrical
length. Theoretically, either method could be used with cables of
any physical length provided the testing frequency were chosen
properl3\ The specific measuring schemes described here are well
adapted to practical application, however.
Short Cable Method ^
When the bridge measurement is made with the distant ends of
wires 1,2,3 and 4 open, as shown in Fig, 13, the impedance of wire 1
to 3-4 is compared to the impedance of wire 2 to 3-4. Assume a
r-x r-x X X
2 2 F 2 2
=rCi
4=C:
=rC,
-03-4
4=C;
p t I LzJL LiL JL J_
2 2 2 2
-o2
Fig. 15 — Equivalent circuit — .short cable method for locating a series resistance
unbalance.
testing current of sufficiently low frequency that the wires are elec-
trically short. Calling the capacitance and the conductor resistance
of the length {T — D) oi each wire, Ci and (r — x), respectively, and of
the length D of each wire, C2 and x, respectively, the bridge circuit of
Fig. 13 is practically equivalent to that of Fig. 15.
The impedance presented to the bridge terminals by the network
*The short cable method is described briefly in the paper, "Cable Testing," by
E. S. Ritter, loc. cit.
402
BELL SYSTEM TECHNICAL JOURNAL
containing F can be determined by inspection to be:
1
jcoC
2 r
2 ./C0C2
o+^ + tV + tV
2 jwCi 7C0C2
where / is the operator /—I and w is 27r times the testing frequency.
Likewise, the impedance presented to the bridge terminals by the
network containing R is:
1
Z^ = R +
X .jo)C
2 jcoC-i
r 1 1
2 jijoCi jcjoCi
When the bridge is balanced, these two impedances are equal, so that:
- +
2 JC0C2
r 1 1
2 J^^\ J0JL2
= R,+'^^+^"^
2 JC0C2 J
This equation reduces to:
r 1 1
2 ./'^L-i JC0C2
1
+ T
1
2 JwCi 7a;C2
r 1 1
--j--^^^ h ^
2 JojCi j(joC2
2^
For a testing current of relatively low frequency the capacitive
reactances, l/jcoCi and I/JC0C2, are much larger than the resistances,
r and F, and the above equation can be written as follows, the symbol
= being used to denote "is practically equal to":
' + '
jooCi JWC2
'^0
F
1 2^
jwC: J Rq '
C2
Ci + C2
Since d is proportional to the length D and (Ci + Co) to the total
length, T:
/Ro ^ D
I F ' T
When the bridge is balanced to the value R,, with the distant ends
of wires 1, 2, 3 and 4 connected together, the amount of unbalance
between wires 1 and 2 is measured. Assuming that F is the only
unbalance present, and that the conductor resistances of wires 1 and 2
BRIDGE METHODS FOR LOCATING RESISTANCE FAULTS 403
are eciual :
and therefore:
Re- F
D = t/^ . (11)
Applicntion: It will be clear from the above theory that Formula
(11) will give accurate results only if the following requirements are
met:
1. The resistance, F, must be the only unbalance on the wires.
2. The resistance of the unbalance must remain constant throughout
a set of measurements to determine i?o and Re.
3. The conductor resistances of wires 1 and 2 must be equal.
4. The capacitive reactances of wires 1 and 2 to 3-4 must be large
as compared to the conductor resistances of the wires and the
fault resistance.
5. Capacitance unbalances of wires 1 and 2 to 3-4 must be negligible.
In general, the short cable method is suitable for locating, with a
fair degree of accuracy, series resistance unbalances ranging from a
few ohms to several hundred ohms on non-loaded cable not exceeding
three or four miles in length. In cases of unbalances of only a few
ohms resistance, however, it is essential that the wires of the faulty
quad be very well balanced in conductor resistance; and the bridge
rheostat should be variable in steps of 0.1 ohm. Usually, best results
are secured when measurements are made from the cable end nearer
the fault.
The bridge voltage used should be as small as practicable in order
to minimize changes in fault resistance. A sufficient number of separ-
ate determinations of the location should be made to insure that con-
sistent results are being secured.
The measurement with the distant ends of wires 1, 2, 3 and 4 con-
nected together is made merely to obtain the actual value of fault
resistance. The value of fault resistance can be obtained instead by
a d.-c. Varley measurement, if desired. If this is done, however,
arrangements should be made so that the bridge connections can be
changed rapidly, as it is desirable to make measurements of R^ and
Re in quick succession to avoid errors due to changing fault resistance.
The short cable method is applicable to paired cable as well as to
quadded cable. In the case of paired cable, ground may be substi-
tuted for wires 3-4, and measurements made of impedance to ground
rather than of impedance between wires. Usually in these circum-
stances, however, the bridge cannot be balanced very sharply.
404
BELL SYSTEM TECHNICAL JOURNAL
Long Cable Method ^
Referring to Fig. 14, assume that the wires under test are non-
loaded and that a testing current of very low frequency is used so
that the wires are electrically short. Calling the capacitance and the
2 2 "^ 2 2
Fig. 16 — -First equivalent circuit — long cable method for locating a series
resistance unbalance.
conductor resistance of the length {T — D) of each wire, Ci and (r — x),
respectively, and of the length D of each wire, C2 and x, respectively,
the bridge circuit of Fig. 14 is practically equivalent to that of Fig. 16.
When the bridge is balanced so that there is no current through the
detector, the impedance Zi looking into the upper branch of the net-
£zA 111 rtx r+x
2 2 F 2 2
Fig. 17 — Second equivalent circuit — long cable method for locating a series
resistance unbalance.
work must be equal to the impedance Z2 looking into the lower branch.
At the balance point the bridge circuit is practically equivalent to
that shown in Fig. 17, in which the network up to the point of fault,
as seen from the bridge terminals of the lower branch, is replaced by
a single resistance-capitance network.
' Credit for the long cable method is given to Capt. F. Reid in the paper, "Cable
Testing," by E. S. Ritter, loc. cit.
BRIDGE METHODS FOR LOCATING RESISTANCE FAULTS 405
The network of Fig. 17 can be replaced by the equivalent network
of Fig. 18. The values of the impedances //, k and p of Fig. 18 are:
h='-^-\-
jcoC:
(r+ F)
1
+
1
jcoCi jw{2Ci + Ci)
+ r-\- F
k =
1
1
JC0(2C2 + Ci)
+
1
jcoCi Jw(2C2 + Ci)
+ r + F
^ = L+f + i?o +
1
jw(2C2 + Ci)
(r + F)
+ .
1
JCoCl 7C0(2C2 + Cl)
+ r+ F
AAA/WW^
k
M/WWV*-
-WWVW
P
Fig. 18— Third equivalent circuit— long cable method for locating a series
resistance unbalance.
It is evident from inspection of Fig. 18 that if h equals p the net-
work is balanced so that there is no current through the detector.
Equating the values of h and p, and solving gives:
F
" 1
jcoCi
1 1
+-;
" 1
jcoCi
1
joiilCi + Ci) _
jw{2C2 + Ci) _
1 ,
1
1 771
+
jcoCi JCC{2C2 + Cl)
-j- r -\- F
X
F
If the capacitive reactances of the wires are very high compared to
the conductor resistances and the fault resistance, this last equation
can be reduced to:
Ro
F
a
+
Cl + C2 ' F
C2
Ci4- C2
X
J'
406 BELL SYSTEM TECHNICAL JOURNAL
and since, for a testing current of very low frequency, C2 and x are
proportional to D, while (Ci + C2) and r are proportional to T:
a
and we may write;
C, + G
F ' T
= .T,
\\'hen the bridge is balanced to the value R, with wires 1, 2, 3 and 4
connected together at the distant end, the amount of unbalance
between wires 1 and 2 is measured. Assuming that F is the only
unbalance present, and that the conductor resistances of wires 1 and
2 are equal :
Rr.^F
and therefore;
D = ^T. (12)
Application: The same general requirements set down for the short
cable method must be met to secure accurate results with the long
cable method. While Formula (12) has been developed specifically
for non-loaded cable, it is clear that it applies also to loaded cable,
provided the effective series impedances of the wires, including the
loading coils, are very low compared to the effective shunt impedances
of the wires. A testing frequency of three or four cycles per second
is sufficiently low to satisfy this requirement on telephone cables up
to a repeater section in length. If, however, the cable is only a few
miles in length, the effective sensitivity of the bridge may be too low
for satisfactory results.
In general, the long cable method is suitable for locating, with
reasonable accuracy, series resistance unbalances ranging from about
10 ohms to several thousand ohms. A well insulated bridge and a
fairly sensitive galvanometer are desirable, especially when workmg
with faults of low resistance.
An essential requisite for accurate results is that the resistance of
the fault remain constant while a set of measurements to determine
Ro and Re is being made. In the application of the method, therefore,
the bridge voltage used should be as low as practicable. Bridge volt-
ages of, say, 100 volts for measuring Rq and six volts or less for measur-
ing Re are usually satisfactory. In this connection it can be pointed out
that if measurements Roi and R02 are made from the two ends of the
cable it is unnecessary to measure Re since {Roi + ^^02) will equal F
BRIDGE METHODS FOR LOCATING RESISTANCE FAULTS 407
and Formula (12) can then be written:
D = T
Rqi + -Ro
In cases where the fault resistance appears to be affected appreciably
by the testing current this scheme of measuring may be found desirable.
It has been found that, when a battery and manually operated
battery reversing switch are used and the balance point of the bridge
is determined by observing the galvanometer kicks as the battery is
reversed, the action of the galvanometer is somewhat as follows: For
settings appreciably below the balance point the galvanometer kicks
are definitely in one direction while for settings which are too high the
kicks are definitely in the opposite direction (assuming, of course, that
the polarity of the battery is taken into account). When the rheostat
setting is very close to the point of balance but slightly too low, the
galvanometer gives a quick double kick, i.e., the needle moves away
from galvanometer zero, then returns toward zero a short distance
and again moves away from zero. When the rheostat setting is
slightly too high, the galvanometer gives a single kick and then coasts
toward the end of the scale. The balance point of the bridge is where
the transition from double to single kick occurs.
When the value of Rq is low a rheostat variable in steps of 0.1 ohm
may be necessary if the transition point is to be accurately obtained.
Seasoned judgment is an essential adjunct to a knowledge of theory
in the practical application of fault locating methods. This is espe-
cially true in the case of methods such as those discussed here, with
which accurate results cannot be secured unless the fault resistances
remain constant in value while a set of measurements to determine
location is being made. Experience has indicated that cable faults of
the types discussed are apt to be inconstant in resistance. Great
care must be exercised, therefore, in interpreting the results of meas-
urements. It is very important to make a sufficient number of separ-
ate sets of measurements to insure that consistent data are being
obtained.
Mutual Impedance of Grounded Wires Lying on the
Surface of the Earth *
By RONALD M. FOSTER
This paper presents a formula for the mutual impedance between two
insulated wires of negligible diameter lying on the surface of the earth and
grounded at their end-points. The formula holds for frequencies which are
not too high to allow all displacement currents to be neglected. For any
two elements dS, ds of the two wires the mutual impedance is obtained
from their direct-current mutual impedance by introducing the complex
factor 2(7?-) -2 [1 - (1 + 70e-'>"'] in the reactance term, 7 being the propaga-
tion constant in the earth, and r the distance between the elements dS
and ds.
THE mutual impedance of grounded circuits may be derived from
certain results obtained by A. Sommerfeld/ who has developed
formulae for the electric and magnetic fields in the earth and in the air
due to horizontal and vertical electric and magnetic antenucC situated
at the surface of the earth. For our present problem we use his formu-
Iffi for the electric field in the earth due to a horizontal electric doublet,
since this doublet may be regarded as a short element dS of a wire of
negligible diameter carrying a finite current. At the end of this
present paper we shall show how the same formula for the mutual
impedance may be obtained directly from first principles.
Sommerfeld uses rectangular coordinates {x, y, z) and the corre-
sponding cylindrical coordinates {r, (j), z), the surface of the earth,
assumed flat, being the xy plane, and the s axis extending upward into
the air. The doublet is at the origin, and its axis along the x axis.
Then the components of the Hertzian vector - in the earth (s < 0) from
which the electric field is determined are ^
(1) n.= C*4£:^.w.-rv„
(2) II, = 0,
* Presented by title at the Eugene, Oregon meeting of the American IMathemat-
ical Society, June 20, 1930, as "Mutual Impedances of Grounded Circuits."
lA Sommerfeld, " Uber die Ausbreitung der Wellen in der drahtlosen Tele-
graphic," Aunalen der Physik, (4), 81, 1135-1153 (December 1926). This paper is a
summary and an extension of earlier work by Sommerfeld and von Hoerschelmann,
references to which will be found in the paper. _
2H. Abraham and A. Foppl, "Theorie der Elektrizitiit," 5th ed., Leipzig and
Berlin, 1918; Vol. I, § 79, page 331. . . ,
a A. Sommerfeld, loc. cil., pages 1145 and 1146, introducing the constant factor
defined on page 1152.
408
MUTUAL IMPEDANCE OF GROUNDED WIRES 409
(3) n. = C(A'^ - ^0-^) ^ COS J^ -Ijrp. e'^^Vdp,
where the time factor e'^"' is omitted throughout. Jo is the Bessel
function of order zero, and the constants k and ko are the propagation
constants in the earth and in the air for plane waves varying with the
time as e"*"". Their values in Heaviside units are given by Sommerfeld
as
(4) A'- = - (ew- + -/o-oj), ko- = - eoco-,
where e and eo are the dielectric constants of the earth and of the air,
respectively, a is the conductivity of the earth, assumed uniform, and
c is the velocity of light. In both media the permeability is taken as
unity. Also
(5) N = k-<p' - ko- + ko-^p- ~ k\
(6) N' = Vp- - ko' + Vp^ - k\
and C is a constant measuring the electric moment of the doublet.
We now replace the doublet by a short element of wire dS carrying a
current 7e*"', and at the same time we assume that e and eo are both
negligible, so that all displacement currents are neglected. This is a
simplification which is ordinarily made as a first approximation at
power frequencies for the shorter transmission lines. Then, introduc-
ing c.g.s. electromagnetic units, in which the conductivity of the
earth is X, and noting that we have changed the sign of w, formulae
(4)-(6) become
(7) k- = — i4Tr\co = — 7^,
(8) ^0^ = 0,
(9) A^ = - TP,
(10) iV' = P + Vp^ + 7-,
and the constant C is such that
r^h 2 1
(11) —T^ = - X current X effective length of doublet
^ Ids
2w\'
410 BELL SYSTEM TECHNICAL JOURNAL
Substituting from (7) (11) in (l)-(3) we have, tlierefore,
IdS f ff'P , d'Q , d'Q
l-n-'Ky- \dz- dx'-dz ' dy'd
(13) II, = 0.
. , Ids d r
Ids d r^ Mrp)
p -\- ^p' -\- 7-
Ids / c3-P d'Q
e'^^'+^'dp
2Tr\y-\dxdz dxdz"
where
(15) P= r J,{rpy^'''-+~''-£l
Jo Vp- +
7"
and
(16)
= r Mrp)c^''^^^-J^
Jo Vp- + y-
= 7o[|7(/^ + c)]A'o[^7(^ - s)].
with i?- = r- + S-.
The integral P is well known/ while Q is evaluated by a suitable
transformation of a Fourier integral.^ /o(s) = /o('-) and ivo(s)
= ^iriHo'-^^iiz) are the Bessel functions of the first and second kinds
for imaginary arguments as defined by G. N. Watson.^ In reducing
n^ to this form we use the differential equation ^ for Jq to obtain the
relation
{:^^ -\- i^) Mrp) +pVo(/'p) = 0.
The components of the electric force in the earth are obtained from
11 by the formula
(17) E = graddivn - 7TI,
â– * See e.g. H. Bateman, "Electrical and Optical Wave-Motion," Cambridge, 1915,
page 72; or G. N. Watson, "Theory of Bessel Functions," Cambridge, 1922, page 416,
formula (2) of § 13.47, with // = and u = L
"* G. A. Campbell, "The Practical Application of the P'ourier Integral," Bell
System Technical Journal, 7, 639-707; using pair 936 of Table 1, with a = 5, substitut-
ing X- for (g- — 4) in the integral of G, and generalizing the resulting integral to in-
clude complex quantities.
" G. N. Watson, op. cit., pages 77, 78.
' G. N. Watson, op. cit., page 19, formula (1") of § 2.13.
MUTUAL IMPEDANCE OF GROUNDED WIRES
411
and we thus obtain Ex, Ey, E^ in the compact form
(18)
_ „ „, Ids/ ah
(£.,£„,£.) = 2"-, (-.-^
cvq_
d'-p
(VQ
rV-P
dz" ' dxd \'dz ' dxd:
where P and Q are given by (15) and (16). In deriving this form we
use the fact that Q satisfies the wave equation
X'^ ay- az-
At the surface of the earth {z = 0) the electric force takes the simple
form
(19) (£., E.) = ^
d\~ \ r
1 + ir
cT
d.Vfh' \ r
where we have used the expressions for the derivatives ^ of the Bessel
functions, lo'iz) = Ii(z), Kq{z) = — Ki{z), and also the identity^
h{z)K,{z) + h{,z)K,{z) = 1/s.
The mutual impedance dZi^ between two infinitesimal elements dS
and ds is now written down as the ratio of the resulting electric force in
one element to the current in the other, with sign reversed:
(20) dZ^i
dSds
IttX
dSds
27rX
cos e -T—; - — COS e r. C~
d \~ \ r / r
sm e
d-
dxdy \ r
3 sin $ sin </> — cos e cos e ., , ^ ^
where $ and are the angles which the elements dS and ds make with
r, and e == $ — is the angle they make with each other.
Integration over the two wires 5 and 5 gives a general formula for
the mutual impedance of grounded wires lying on the surface of the
earth:
(^') ^'-2^//l7ls(-.)+^t.-(. + ..).-"]}^.
Sds
= //[
J ^ / 1
27rX ' dSds
+ fco ^1 ^, [1 - (1 + 7r)e-] } ] dSds.
* G. N. Watson, op. cit., page 79, formula (7) of § 3.71.
' G. N. Watson, op. cit., page 80, formula (20) of § 3.71, with v = <d.
412
The factor
(22)
BELL SYSTEM TECHNICAL JOURNAL
-,[1 - (1 +7'')6'-"^]
hrf
approaches unity as w approaches zero, and Zio then agrees with the
5 6
VALUES OF r-'
¥\g. 1 — Real and imaginary parts of the complex factor,
2
(7^)-
,[!-(!+ yr)e-^'\
plotted as functions of r' = |t''I = (47rXw)''-r.
MUTUAL IMPEDANCE OF GROUNDED WIRES 413
direct-current mutual impedance as given by G. A. Campbell. i°
Introducing this factor, which is a function of yr only, into the re-
actance term for the direct-current mutual impedance between two
elements dS and ds gives the general expression for their mutual im-
pedance corresponding to the propagation constant 7. It is interesting
also to determine, for any given value of 7, the variation of the factor
(22) for increasing values of r. This is shown very clearly in Fig. 1,
where the real and imaginary parts of (22) are plotted for increasing
values of / = [7^! = (47rXa))^/V. The real part, we note, decreases
rapidly from the initial value unity as r' increases, while the imaginary
part is always negative, decreasing from zero to a minimum value
(approximately — 0.3 for r' = 1.5) and then increasing towards zero,
although it does not approach zero so rapidly as the real part does.
The first three terms in the expansion of Z12 for low frequencies
are given by
(23) Z^.. = ^l^-^--l+^)+io.Ns.
lirX \Aa Ab Ba Bb
-r (1 - i)^(S ttXoo'Y'- A B ab cos d -\- •••.
where Nss is the mutual Neumann integral between the two wires S
and 5 of arbitrary form but with end-points A, B and a, b respectively;
d is the angle between the straight lines AB and ab. The first two
terms in this expansion are precisely the direct-current mutual im-
pedance as given by G. A. Campbell.
The first term in the expansion of Zn for a long straight wire S and
any wire s located near the midpoint of 5 is
(24)
/
-T— , — Ai(7.v,
TTAA'- TTAA
COS e ds,
X being the positive distance from ds to 5, and e the angle between ds
and S. Kilz) = - ^tHi^^^iz) is the Bessel function of the second
kind for imaginary argument as defined by G. N. Watson." In ob-
taining (24) from (21) we use the derivative with respect to x of the
integral
-yr
dz ^ Ko{yx),
which is a special case of the integral used above in evaluating Q, with
X assumed positive.
^°G. A. Campbell, "Mutual Impedances of Grounded Circuits," Bell System
Technical Journal, 2, 1-30 (October 1923).
" G. N. Watson, op. cit., page 78.
414 BELL SYSTEM TECLINICAL JOURNAL
The expression in square brackets in (24) is the mutual impedance
gradient parallel to an infinite wire at a positive distance x from the
wire. It agrees with the results published independently by F.
Pollaczek/- J. R. Carson,'^ and G. Haberland/^ eind has been employed
by us to obtain numerical results since 1917. Pollaczek has also in-
vestigated the case of two gounded circuits of finite length. ^^
The mutual impedance dZx^ between a short grounded circuit dS
and a counterclockwise small loop of area da, on the surface of the
earth, is given by the formula
,^r-^ 7-7 dSda sin </) p- /-> i ^ i o on ^^,-1
(25) dZ^o = -^^ • —^l^ - (3 + ^ir + 7-r)e ^'■],
where (/> is the angle which dS makes with r, the line from da to dS.
This may be obtained from Sommerfeld's formulae for the horizontal
electric force due to a vertical magnetic antenna, or it may be obtained
by an application of Stokes's theorem to formula (20) above.
By a further application of Stokes's theorem we may obtain the
mutual impedance between two counterclockwise small loops dA and
da, namely,
(26) dZu = ^ • 1 [(9 + 97r + 47^'^ + yr')^-^'- - 9].
This result might also be derived from Sommerfeld's formula for the
vertical magnetic force due to a vertical magnetic antenna.
We shall now indicate briefly how the same value of E as given in
(18) above may be obtained directly, though more laboriously, from
first principles. In this method we start from the fundamental
solution ^^
(27) u = e'-^+'»!/+"V'
of the wave equation
^ ^ dx~ ay- oz-
12 F. Pollaczek, " Uber das Feld einer unendlich langen wechselstromdurchflossenen
Einfachleitung," Elektnsche Nachrichten-techuik, 3, 339-359 (September 1926).
'3 J. R. Carson, "Wave Propagation in Overhead Wires with Ground Return,"
Bell System Technical Journal, 5, 539-554 (October 1926).
"G. Haberland, "Theorie der Leitung von Wechselstrom durch die Erde,"
Zeitschrift fur angeimndte Mathematik und Mechanik, 6, 366-379 (October 1926).
'5 F". Pollaczek, "(iegenseitige Induktion zwischen Wechselstromfreiieitungen von
endlicher Lange," Annalen der Physik. (4), 87, 965-999 (December 1928). His as-
sumptions regarding conditions at the ground connections seem to depart considerably
from the conditions assumed in the present paper, and moreover his results are not
expressed in convenient form for direct comparison with the formula given above
for Z\i.
"H. Bateman, op. cit.,% 4, pages 6, 7; § 11, page 26.
MUTUAL IMPEDANCE OF GROUNDED WIRES 415
which is satisfied by the electric force in the earth; 7 = {iAivX^Y''- is
the propagation constant for plane waves which vary with the time as
f*"'. The parameters /, m, n satisfy the relation
(29) /- + ni- + 7/2 _ ^2 = Q^
In the air, the same equations hold, but with the propagation constant
7 equal to zero, and we note that the solution in the air must be chosen
to vanish at an infinite height, while in the earth the solution must
vanish at an infinite depth.
For convenience in this method we start with a short straight wire of
length 2a lying along the x axis, later allowing a to approach zero.
Thus we suppose that the current /e*"' enters the earth at the point
(a, 0, 0) and leaves it at the point (-a, 0, 0). The factor e^"' will be
omitted, however, throughout the following work. The current flow
in this system is symmetrical with respect to the vertical plane through
the wire, the xz plane, and is also symmetrical, but with sign reversed,
with respect to the vertical plane normal to the wire at its midpoint,
the yz plane. Then if we replace the three parameters /, w, n of (27)
by two independent parameters m. J^. such that
(30) / = ± /m, m = ± iv, n = ± Vm' + f' + 7',
formula (29) is identically satisfied, and we can then replace the four
solutions e"^^'^'^'"" by their corresponding expressions in terms of sines
and cosines, namely,
sin XjjL sin yp, sin Xfx cos yv, cos xii sin yv, cos .y/x cos yv.
The above considerations of symmetry will eliminate, for each com-
ponent of the electric force, all but one of these forms. With the re-
maining solution as a basis we build up, by means of the Fourier in-
tegral, a general expression for any possible steady harmonic oscilla-
tion. Hence we may write down the general solutions for the total
electric force in the earth (s < 0), as follows.
(31) £x = F,{iJL,v)e'^''"-+'"+''' cos Xfx cos yudfxdv,
Ju Jo
I fyil^, J/) 6^^"'+"-+^' sin xn sin yudndv,
- „ Jo
I /^,(m, J/) 6^^"^+"=+^' sin xix cos yv dfxdu,
Jo
(32)
(33)
416 BELL SYSTEM TECHNICAL JOURNAL
where the positive sign is chosen in the exponential term containing s
since the solution must vanish at an infinite depth, s being negative in
the earth ; and that value of the radical is taken which has a positive
real part. Fx, Fy, Fz are arbitrary functions of their arguments, to be
determined by the physical conditions of the problem.
In the air (0 < z) we may formulate the corresponding solutions for
the total electric force as
(34)
(35) E
(36)
I I P^ifjL, p)e-^^'''+'''cosxiJLCOi^yvdiJ.(h,
y = i I PyiiJi, v)e-^'''+'"s'in Xfi sin yvdjxdv,
Jti Jo
= I I F.iij., v)c-'^^-'^''^ s\n Xjx COS yvdixdv,
Jo Jo
where the propagation constant is zero in the air; the negative sign is
chosen in the exponential term containing z since the solution must
vanish at an infinite height, z being positive in the air; and Px, Py Pz
are arbitrary functions of their arguments.
To determine these six arbitrary functions we need six independent
relations among them. Two of these relations are obtained by
utilizing the fact that the divergence of the electric force either in the
earth or in the air is equal to zero, that is,
dEx dEy dE,^
dx dy dz
By means of this we obtain from (31)-(33),
(37) - fxF, + vFy + Vm' + j'- + y'F, = 0,
and from (34)-(36),
(38) - fiP, + vPy - ^ix' ± z^-P. = 0.
Since the horizontal components of the electric force are continuous
at the surface of the earth (s = 0) we see that we must also have, from
(31) and (34),
(39) F, = P.,
and from (32) and (35),
(40) Fy - Fy.
MUTUAL IMPEDANCE OF GROUNDED WIRES 417
We may obtain a fifth relation from the fact that the current /
flows through the earth from one grounding; point to the other. To
utiHze this fact let us compute the total current flowing out through
five faces of a rectangular prism in the earth, the sixth face being a
rectangle in the surface of the earth surrounding the grounding point
(a, 0, 0), the prism extending from .r = a — ^ to .r = a + ^, from
y = — 7] to y = Vi and from ;: = — j" to 2 = 0. The components of
the electric force being given by (31)- (33), and X being the conductivity
of the earth, we obtain for this current the expression
(41) - 4X f rV /in aM sing, sin ,. ^^^^^^^
Jo Jo 1^"
after simplifying by means of the divergence condition (37). This
current flowing out through the prism is / if the face in the surface of
the earth includes only the one grounding point (a, 0, 0), but is zero
if it includes both grounding points; that is, the above integral (41)
equals / if ^ < 2a. but equals zero if 2a < $, for any positive value of r?.
It is readily verified that the Fourier integral
(42) ^ r r^'"^^A/sin?Msin..^^^^^
has the desired properties. Accordingly, we must have
2/
(43) F^= — ^sina/i.
TT-A
To obtain the one additional relation which is needed, we make use
of the fact that the current / flows through the straight wire from one
grounding point to the other. Let us integrate the magnetic force
around a rectangle in a plane perpendicular to the wire, that is,
perpendicular to the x axis, the rectangle extending from y — — r]
to y = 7} and from s = — f to s = f , the path of integration being
taken in the clockwise direction looking along the positive direction of
the X axis, and then equate this integral to 47r times the total current
threading the rectangle. The components of the magnetic force
which we need, Hy and Hz, are found from the fact that curl E= —iuH,
that is,
(44) io:Hy
(45) iccH,
dE,
dE,
dx
dz
dE.
dE,
dy
dx
418 BELL SYSTEM TECHNICAL JOURNAL
where the £'s are given by (31)-(33) for s < and by (34)-(36) for
< 2. We now subtract from this integral 47r times the current in
the earth which threads the rectangle, this quantity being found by the
appropriate integration of E^, as given by (31), over that portion of the
area of the rectangle which lies below the surface of the earth. As a
final result w^e obtain the expression
(46) - r C-i- Vm'^ + V- + y-'F, + ^iF. - Vm'-^ + v'Pr - ixP.)
'^ Jo Jo
X COS xn sin r]v djxdv,
after simplifying by means of the divergence conditions (37) and (38).
The net current threading the rectangle, after subtracting the current
in the earth, is / if the rectangle is situated between the two grounding
points, but is zero if it is outside them; that is, the above integral (46)
equals 4x7 if \x\< a, but equals zero if a < |:x;j , for any positive value
of Tj. It is readily verified that the Fourier integral
167 r°° r" sin an cos xjj. sin 171' , ,
~^ Jo Jo ^'^
has the desired properties. Accordingly we must have
(47) - Vm- + V- + tF, + ixF, - VmM^'/'x - iJ^Pz
_ 87w/ sin a IX
TV jX
We can now solve equations (37)-(40), (43), and (47) for the six
arbitrary functions, obtaining
(48) F. = P. = -^
_ \M- + V- + 7-
_ mVm' + v'~
sm an,
(49) Fy= Py= ^- . /^ ., sin a IX,
21 .
(43) Fz= ^sin Gju,
(50) P. = ^^^ ^^' + »-' + ^'^ sin an.
Substituting these values in equations (31)-(33) and letting a
approach zero such that 2a = dS, we find, for the electric force in the
MUTUAL IMPEDANCE OF GROUNDED WIRES 419
earth.
(51) £. = ^ r r r^^i= - <^^ + ^ + A .^v:
^"^ Jo Jo L Vm' + i'' J
jj2-)-v2-f--y2
X COS x/i cos yv dfidp,
(52) Ey = ^^ r r . '-"' eW/l^+i:i+^ sin x^ sin 3,^, ^^,/^^
^"-^ Jo Jo Vm- + i^'
(53) E,= --^l i fxi^'^'^'+^'+y' sin xfi cos yudndu.
These are precisely the values found by the former method, for the
integrals P and Q may be expressed as double integrals by substituting
for Jo{rp) the integral expression given by the formula ^"^
I 2 f^'^
(54) Jo{ryJ fx- + v-} — — | cos(r/i cos d) cos {rp sin ^)^^,
^Jo
and introducing rectangular coordinates in place of r, 6. These inte-
grals may, therefore, be written in the equivalent forms,
• 00 ^00 g2VM2+l'2+72
(55) P = — \ I , " COS XjjL cosyy dfxdp,
""Jo Jo - â– " â– ^
2 /'•oo /-.oo ^Z-\Jn2+v2+yi
(56) <2 = — I I cos xiJL cos ypdfjidp,
^Jo Jo Vm' + J^'Vm' + J'' + t'
and comparison with (51)-(53) again leads to the values
(18) (E E E)=^(-^^-'^ -^^ ^)
where P and Q are evaluated in (15) and (16). Thus the mutual im-
pedance formula presented in this paper may be derived directly from
first principles, without reference to the work of Sommerfeld.
I am greatly indebted to my colleague. Dr. Marion C. Gray, for
putting into its present form the derivation of my formula from
Sommerfeld 's results.
" G. N. Watson, op. cit., page 21, formula (1) of § 2.21.
Transients in Grounded Wires Lying on the
Earth's Surface*
By JOHN RIORDAN
Voltages during transient conditions in a grounded wire lying on the
earth's surface due to current in a second grounded wire also on the earth's
surface are formulated for types of transient currents ordinarily obtained
in a.-c. and d.-c. circuits. The fundamental formula is for voltage due to
a unit step current, that is, a current zero for time less than zero, and unity
for time greater than zero; curves are given for the function determining
this voltage for a wide range of values of its two parameters. The for-
mulas for other types of currents are not well adapted for numerical com-
putation, which should be more conveniently carried out by numerical
integration using the above curves.
I
A FORMULA for the mutual impedance of grounded wires lying
on the earth's surface has recently been published by R. M.
Foster.' The object of the present paper is to derive formulas for the
voltages during transient conditions in one such grounded wire due to
current in a second for types of transient currents ordinarily obtained
in a.-c. and d.-c. circuits, and particularly for the voltage due to unit
step current, zero for time less than zero, unity for time greater
than zero.
The voltage due to unit step current is expressed in closed form for
straight parallel wires; closed form expressions have not been obtained
for straight parallel wires for the exponential forms of current for
a.-c. and d.-c. transients. While the integrals might be evaluated
numerically, or transformed to asymptotic expressions, it appears
more desirable in practical calculation to use the curves given for the
unit step voltage directly; a single integration is necessary to find the
voltage for current of arbitrary wave form, from the unit step result.
The fundamental physical assumptions upon which the steady-state
formula is based are as follows: The surface of the earth is assumed
flat, the earth semi-infinite in extent, of uniform conductivity X, unit
"â– A brief report of the results in this paper was given at the Summer Convention
of the American Institute of Electrical Engineers, Toronto, Ontario, Canada, June
23-27, 1930, in Discussion of "Mutual Impedances of Ground Return Circuits — •
Some Experimental Studies," by A. E. Bowen and C. L. Gilkeson; A. I. E. E. Trans.,
Oct. 1930.
' R. M. Foster: "Mutual Impedances of Grounded Circuits" (Abstract), BiiUetiu
of the American Mathematical Society, May, 1930, pp. 367-36iS; "Mutual Impedance
of Grounded Wires Lying on the Surface of the Earth," Bell Svstem Technical Journal,
Julv, 1931.
420
TRANSIENTS IN GROUNDED WIRES 421
permeability and negligible dielectric constant. The air above the
earth is of zero conductivity, unit permeability, and negligible dielectric
constant. Because of the assumption of negligible dielectric constant,
the formulas for voltages during transient conditions do not hold
strictly for small values of the time, that is, during the initial stages of
the transient. The wires are of negligible diameter, lying on the
surface of the earth, and insulated from it except at the ends, where
there is point contact.
In using the steady-state solution as the basis of transient solutions,
the Heaviside operational calculus is employe d af ter replacing 7co,
where co = lirf is the radian frequency and i = V— 1, by p = d/dt, the
time differentiator, since (rf"/W/") (exp icot) = (/w)" exp ioot, where n is
integral.
II
The mutual impedance of grounded wires lying on the surface of
the earth and insulated from it except at the ends is given by the
following formula: ^
The integration is extended over the two wires S and s, having
arbitrary paths, r and e are the distance and angle, respectively, be-
tween differential elements dS and ds, and 7 = {'iTrXiooY'^; X is the
ground conductivity and w = 27r/ is the radian frequency.
Replacing ico by p = d/dt in 7, the resulting forms to b e evaluated
are exp ( — a>[p) and ^ip exp ( — aVp) where a = rV-ivrX. The first
of these is known and, following Heaviside,^ may be developed as
follows.
Expressing the exponential in series form:
exp(- ay^p) = 1 - ^^^+_i^_-^^+ ....
Integral powers of p are neglected, since (omitting the discontinuity
at / = 0) the operand is unity and the derivative of a constant is
zero. Then:
exp (— a4p) = 1 — a\^
<iP a-^
^ i\ ^ 51 ^
The bracketed terms may now be assumed to operate on y[p = (x/) "^
2 Foster, loc. cit.
3 Heaviside: "Electromagnetic Theory," Vol. II, pp. 49-51, equations (4) and (12).
a
422 BELL SYSTEM TECHNICAL JOURNAL
and, if />" is replaced by (I"ldt",
[ 3.rl!\4// ^5.v2!V4//
= 1 -erf-^,
2V/
since the term in brackets with its accompanying multiplier is the
absolutely convergent expansion of the error function (erf) ;
2 C^
erf (n) = -p | exp ( — z^)dz.
Vtt Jo
The result may also be established either by use of an integral
equation •* or the Fourier integral; it is given as pair 803, Table I, in
tables published by G. A. Campbell.^ In the present use of the tables,
for unit step current, the mate of F{p)lp, where F{p) is a function of p
to be evaluated, is taken since the unit step function is expressed by
p-^ (pair 415).
The second operational form required may be derived from the
first by differentiating with respect to a, since {dlda)F{p) = {dfda)f{t)
where F{p) and /(/) are corresponding functions of p and /. Thus,
ayip exp ( - ayjp) = -p exp f - ^ j ,
since
^^erf IHOI =^^'(0 exp | - IHDJ } '
The unit step voltage may now be expressed, by substitution of
these results, by the following formula:
2r^/-exp( --y-
dSds. (1)
In equation (1), as in the steady-state formula from which it is
derived, the wires are unrestricted in path or length on the surface of
* J. R. Carson: " Electric Circuit Theory and The Operational Calculus," McGraw
Hill Co., 1926, p. 19, eq. 29.
* "The Practical Application of the Fourier Integral," Bell System Technical
Journal, October, 1928.
TRANSIENTS IN GROUNDED WIRES 423
the earth. The formula for straight parallel wires, wire 5" extending
along the s axis from — a to + a, and wire 5 from Si to z^ at distance
X from it, is obtained by double integration between these limits with
^2 = x^ + {S - 5)2, cos e = 1.
The result of integrating once, with respect to S, is:
V.v2 + (a - 5)2 ^Jx' + (a + ^)' J
+ 0(5 + a) - 0(5 - a) I (Is, (2
where
(t>{n) = " -erf ( V.v' + u~ J^
x-^{x^ + It-) \ ^ ^
1 / 7rX.r2\ ./ [^
where u is to be replaced by 5 + a and 5 — a in equation (2).
Equation (2) is checked as follows. In the first term substitute
limits after removing differentiation and integration with respect to
S, which cancel each other. In the second term integrate by parts:
/ ,,+ '-.,J. «fVf[.v'+(^-^)-]'^-^
â– ^"' ierfJ^[.v^ + (5--0=]
The integral coming from this operation combines with the remaining
term to give :
-2^/lexp{-^[..' + (5-.n}</5,
which can be simplified in terms of the error function to the form in
equation (2).
Integration from Si to z^ gives the result:
Fi2(0 = tV ['/'(S2 + a) - '/'(S2 - a) - '/'(si + g) + H^i - «)], (^^)
ZTTaX
where
^(«) = - w+i? + ""^ "' \ " + ^''^
7/. / ttX-V^ \ / /ttX
424
BELL SYSTEM TECHNICAL JOURNAL
As before, u is to be replaced in the equation by tiie functional
arguments, which are the four sums of the s-coordinates of position.
The factor x in \p{u) is introduced to make it a function of two
parameters, ux~^ and ttXx^/"'; the result of integration is x~^\}/{u).
The result has the dimensions of abohms when all quantities are in
electromagnetic c.g.s. units.
To check equation (3) notice that the integration of the first term
of equation (2) is effected by removal of differentiation and integra-
tion signs, and substitution of limits; its contribution is identical with
the d.-c. mutual resistance.® The integration of 0(w) may be effected
by integrating the first term by parts and employing the indefinite
integral:
erf (ax)dx = x erf (ax) H pcxp (— a-x"^) -f const.
/'
The result is checked by differentiating, that is, by the relation:
du
X ^\p{ii) +
1
V.v^ + it^ -
For large values of ii,
1 — exp
= 0(")-
rX.r^
/
smce
erf (± =^) = ± 1,
so that for a = co the unit step voltage approaches the limit:
7rX.%'"
VM =
7rX.V-
/
ttXx^
1 — exp
1 - exp -
/
7rX.r^
t
where / = S2 — -i is the length of the second wire.
This result is in agreement with a result published by F. Ollendorff,
Elektrische Nachrichten — Technik, October, 1930, eq. (26), and by L. C.
Peterson, Bell System Technical Journal, October, 1930, equation (5).
The case of collinear straight wires is obtained by taking the limit
^c = 0, which gives
lim X ^\p(7i) = -
X=Q u
- 1 +
/I
V2
ttXh-
t
erf
ttX
U A
\ t
w-'f(«).
exp
r)]
This result involves the evaluation of an indeterminate form.
^ G. A. Campbell: "Mutual Impedances of Grounded Circuits," Bell System
Technical Journal, October, 192.^, eq. (3), p. 5.
TRANSIENTS IN GROUNDED WIRES
425
VALUES OF UX-I
pig_ i_^(m) for the range in which xp{u) < 1, < ux'^ ^ 10.
1.0
0.9
0.8
0.7
1
1
/
1
/
/
/
/
/
20/
40/
60
/
80 /
100/
/
/
/
/
/
/
/
/
/
/
/
y
y
/
/
/
150
^
/ /
/
X
^
^
0.4
0.3
/
^
y
-^
200
^
//
^
y
^
_^
0.1
^
t^
^
^
^^
^
^^
■—
1000
'
,^
00
r t
Tr7vx2
-0 ?
40 50 60
VALUES OF UX~I
80
Fig. 2—^p{u) for the range in which \Piu) < 1, 10 < ux'^ < 100.
Curves for xl^iu) as a function of ux-^ with //(-n-Xx-) as parameter of
the curve families are shown on Figures 1, 2, and 3. The range
yp{u) < 1, is shown on Figures 1 and 2 for ux~^ ^10 and 100, respec-
tively; both figures cover the entire range of //(ttXx") in the intervals.
The remaining range \l/{u) > 1 is shown on Figure 3. For the greater
part of the range on Figure 3 the function is determined by its limiting
form for ux~^ large, that is, by the equation
^(«) = HX~^
1 - exp -
ir\x'
TRANSIENTS IN GROUNDED WIRES
427
^ —
\
â– v
.N.
S>i
s
V s
s
\
s
\
S
^ '^
s
N
\
\
\
•"
N
\
\
\
^\
"^
\
\
\
\ ^
s
\
^
^
\
\
N^
\
\
\
\
\
s
\
\
>^
^
s
N
N
\
\
\,
\
'V
s
s^
X \ ^
s.
s
, \
V
"^
V \^
\
s
's '
N
\ \ V
\
S,
\
\
> s
^
\ \
s
\
s
\^
\
\
\
^
\
S
\
\,
\^
\
s ^
s
\
^
\
\
\
s
\
\
\
\
\^
s
\
\
s
;\
s
"<
\
\
s
\
N^
^
N
^
^
\
\
\
\
N^
\
>•
\
N
\
^
N
\
^
;^
^:
\
\
\
\
\
^'
V
\
\
\
\
\
\
f\
s
v
N ••
â– v \ \
N,
s.
W ^
Sy
s,
(->
\
\
s
s
\ \ s
\,
â– v \^
s
^
o
\
s.
s, ^
\ \
s ^
\,
\ \ ^
s
\,
\ -
S
s
N
\ \ \
N
V
s
\ N
s
s
s
\
\
\
\
\\
s
\
s
\
N. ^
s
s
\
\,
\
\
\
w^
\
\
N
\
\
s
s
\
\
\
. ^
\
\
s \^^
s
\
\
k>
\
V
\
\
\
\
X. 'T
\
\^^
\
\
h^
$^
\
\
\
\
\^
\
§^
\
\
\
^
^
^^^0>^.
s
s
^
k
\
N
f< o ^
s.
N, '
, X X
\,
s
s
\ X
f=
N
s
\
\ \^
^ X ^
^
N,
S, S
\ \
s
\,
\ \
D _j
\
s
\,
s
S
\
s
\ X
n
\
\
\
s
\
\
\
\
D
\
\,
\,
N
\
v^
\
\
>
\
\
\
\
\
;>
\
\
\
\
\
<:
^
^
\
\
\
\
\
\
\
^
^
%^
s
s
s.
S, '•
\ ^
s
\
\.
\,
\ \
\
\,
\ ^
•v\ ^
s
\
W
\
\
s
vX^
\
\
\ \
\
^v \
\
.\
— - — rt
H
o ^
VALUES OF i;;(u)
428
BELL SYSTEM TECHNICAL JOURNAL
or
log \p{u) = log ux-'^ + log
1 - exp -
ttXx^
Thus Figure 3 may be used to indicate the range of applicability of
the limiting form, which is quite large; in this range the unit step
voltage is simplified as shown above.
8
I'
UJ
_l
\
\
\
v
\
\
\
S
s
>
V
^^
L
â– â–
.
-2
VALUES OF
1TAu2
Fig. 4 — The function ^iu), for collinear straight wires; for values below the range
1 , ttXh-
shown f (?<)
+
The function '^{u), for the case of collinear straight wires, is shown
on Fig. 4 for values of the argument //(ttXm^) fi-om 0.1 to 1000; for
small values of the argument, the function is approximately
, , 1 , ttXm"
xXm^
< 0.4
These curves may be employed to obtain voltages due to other
forms of disturbing currents by numerical or mechanical integration
of the following integral : ^
£12(0
/(r)Fi2(/ - r)dT
r) T''i2(r)f/r,
where I{t) is the disturbing current as a function of time.
'J. R. Carson: loc. cit., p. 16, eq. (20) and (20a).
TRANSIENTS IN GROUNDED WIRES 429
III
The equation above may be used to obtain a formula for voltage
due to suddenly applied current exp io^t; or the operational product,
of which it is an expression in terms of /, may be carried out directly
in terms of p. The current is expressed in terms of ^ by:
P
exp ioot = — â–
The second term in ;//(«) is transformed by the operational equivalent
already developed:
erf — p = 1 — exp ( — ayfp) .
The last term in \J/{ii) is not known in closed form in p.
The operational product of exp iuit and the second term is evaluated
by
pl'l - exp (- q;V^)"| _ t> _ P exp ( - a^Jp)
p — 10} p — lb) p — Ico
1
exp loot — -
n — / ^ P — "
exp (;co/ — aV/co) erfc ( — p — yiosf
+ exp {io:t + aV^) erfc ( — p + yuot j
the last term of which is given by pair 819 (with /S = 0) in the tables
referred to. Erfc is the error function complement;
erfc (s) = 1 - erf (2).
The operational product of exp iwt and the last term in ^(11) may
be expressed in integral form by the formula :
^^fu) =
p — iO)
p — tw
= /(/) + io) exp io)t I exp (- iwt)f{t)dt.
Jo
The complete expression for the voltage due to cisoidal current is
as follows:
£12(0 = ttV [*(22 + a) - *(22 -a) - $(si + a) + $(01 - a)], (4)
430
where
BELL SYSTEM TECHNICAL JOURNAL
Vx2 + «2 r
exp
exp (iw/ - tVa^ + ^2) erfc ( A/y (-"^^ + «^) - V/w/
ttX
+ exp (7a;/ + tV.V'' + w2) erfc (^y (.v^ + ^/^j + Vn^
too ex
X
picot I exp ( - iwt - ^^— ^ j erf ( ?/ a/— ) ^/Z-
The integral appearing in <J>(?0 apparently cannot be expressed in
closed form in terms of known functions; for numerical results series
or asymptotic expressions may be derived but it appears more desirable
to employ numerical or mechanical integration using the unit step
voltage since tables or charts of the error function of complex variable
which also appears in <J>(m) are not available.
A useful check on the above formula is obtained by taking the limit
for / = 00 , which gives the steady-state mutual impedance between
straight parallel wires; the result is as follows:
Zi2 = En{t) exp (- iut)
= ttV [^(22 + fl) - ^(S2 - a) - ^(si + a) + ^(2i - a)], (5)
where
^(zO =
-^x^ + n^
x^lx^ + u'^
exp (— y^x"^ + «^)
--f
â– 'â– Jo
exp I — w
Yx^
eri — j= aw
Vx^ + u'^
+
Vx^-f- 7(''
1 — exp ( — 7 V.v' + ir)
y" r r I
7 I exp ( — 7 V-x^ + ii^)dw,
where as before 7^ = 47rXfco.
The third term in ^{ u) approaches the limit given because
erfc (— ^f^^) = 2, erfc (Vi 00) =0; the integral term as given in
the first form of ^{u) has been transformed by the substitution
TRANSIENTS IN GROUNDED U'lRES 431
The first form of ^'(^0 may be checked directly from equation (3)
by introducing ico = p in the operationally equivalent function of p;
the third term of (3) being expressed by the infinite integral:
Fip) = p e-i"f{l)ilt.
Jo
The second form of "^(u) is obtained by separating the d.-c. mutual
resistance term, and transforming the infinite integral as follows:
express the error function in integral form, put y — 7z;/(2Vw) where
y is the variable of integration for the error function, and invert the
order of integration; thus
I exp — w — - — ei I — -;= aw
V^ Jo ' Jo ' \ '' 4w' / Vzl-
exp ( - 7 Vx- + v^)dv.
The infinite integral evaluated in the third line is No. 495 in Peirce's
" Short Table of Integrals," third edition.
The second form of ^(«) may be verified by direct double integra-
tion of the mutual impedance; it agrees with the known result in the
limit for one wire infinite, and, when expanded in powers of y, with
the terms given in the second form for the mutual impedance by R. M.
Foster, loc. cit.
Expressions for voltages due to suddenly applied currents
exp (— kt) sin co/ or 1 — exp (— kt), which are important forms for
a.-c. and d.-c. networks, may be readily obtained from equation (4),
the first by use of the expression:
exp (— kt) sin cot = y. [exp {— kt -\- it^t) + exp {— kt — iwt)~\
and the second by the substitution — ^ = /co and subtraction from
the unit step voltage.
The results attained in this paper depend in appreciable measure
on advice and suggestions received from Mr. R. M. Foster of the
American Telephone and Telegraph Company; I am also appreciative
of the interest and advice of Messrs. K. L. Maurer and H. M. True-
blood of this company.
Developments in the Manufacture of Lead-Covered Paper-
Insulated Telephone Cable *
By JOHN R. SHEA
This paper describes developments in the manufacture of lead covered
paper insulated telephone cable completed during the past three years.
The introduction describes the manner in which cable is used in the telephone
system and briefly outlines the manufacturing processes and equipment as
they existed about three years ago. The new developments are then
treated in considerable detail, the most outstanding of which are the
application of wood pulp insulation direct on the wire instead of spirally
wrapping manila rope ribbon paper; new equipment for vacuum drying and
storing cable in which a large storage room of unique construction is provided
with conditioned air at a relative humidity of .5 per cent at 100° F.; the
central melting of large quantities of lead alloy and its distribution through
piping systems to a number of lead presses; improved and larger sheathing
presses; and precision electrical testing of the finished cable. Most of these
improvements are incorporated in the new Baltimore Cable Plant of the
Western Electric Company.
PAPER-INSULATED lead-covered telephone cable constitutes
approximately 25 per cent of the Bell System telephone plant.
The cost of new telephone cable each year, including installation,
averages $100,000,000. Developments in the process and equip-
ment for its manufacture are numerous and have been a large con-
tributing factor in the establishment of a high standard of service in
the long-distance communication field. The problems involved in
manufacturing engineering are extremely interesting both from an
economic and technical standpoint to the mechanical and the electrical
engineer, the physicist, and the chemist, and the illustrations which
follow contain fundamental engineering principles of use in many lines
of industry.
Before proceeding directly with these problems, a brief outline of
how cable and its associated apparatus function in the long distance
communication field will be of value. After presenting this broad
picture, the bulk of the paper will be devoted to an engineering dis-
cussion of developments in the process and equipment for manu-
facturing cable as illustrated by recent improvements introduced in the
new cable plant of the Western Electric Company at Baltimore and at
the Kearny, New Jersey, and Chicago plants.
* Presented at A. S. M. E. meeting, Cleveland, Ohio, April 13-17, 1931. Published
in abridged form in Mech. Engg., April. 1931.
432
LEAD-COVERED PAPER-INSULATED TELEPHONE CABLE 433
General Information on Use of Cahle
The rapid increase with which cable is being added to the toll plant is
illustrated quite strikingly by Fig. 1 , which shows the present and
proposed increases in cable in comparison with open wire and carrier
1930 1931
YEAR
1933
Fig. 1 — Present and proposed increase in cable in comparison with open wire and
carrier circuits.
circuits.^ The future scope of this expansion is shown by Fig. 2, which
indicates the present and proposed main toll cable routes in the United
States. The exact program on which these cables will be extended
will depend upon how rapidly the business develops; howe\'er, definite
future plans have been outlined to extend the cable to (3maha, Ne-
braska,** and across the continent to San Francisco, thus replacing
and increasing the capacity of existing open wire lines.
^"Recent Developments in Toll Telephone Service" by W. H. Harrison, Jour.
A. L E. £., March, 1930; Bell Telephone Quarterly, April, 1930.
** This cable was completed in ^Iay, 1931.
434
BELL SYSTEM TECHNICAL JOURNAL
U
o
LEAD-COVERED PAPER-INSULATED TELEPHONE CABLE 435
The elements of a typical cable route are illustrated in the New
York to Pittsburgh cable chart shown in Hg. 3. A Pittsburgh call
originating at a subscriber's station, for example, in Yonkers, New
York, passes through the toll board of the local telephone exchange to
the toll center located at Walker Street, New York City. At this
point the connections are completed for the call to Pittsburgh through
the toll cable circuits and repeater stations between the two cities.
The speech currents as they travel along this circuit diminish in
intensity. Loading coils placed along the cable circuit at regular
intervals reduce these losses to a considerable degree but even with
1 VER. 'â– .,
WEST VIRGINIA
Fig. 3 — Typical cable route.
these it is necessary to supply amplifiers (repeaters) ^^ ^ at intervals of
approximately fifty miles to boost the energy level.
The amount of amplification required for intelligible speech varies
with the resistance of the cable conductors which changes with the
temperature. In order to regulate the amount of the amplification to
compensate for these variations, what is known as a pilot wire regulator
is installed at certain repeater points which automatically adjusts the
gain of the repeaters to correct for the changing line losses.
Difficulty is also experienced on long toll lines due to the voice
currents being reflected back to the speaker. To prevent this, a
device is provided which automatically short circuits one side of the
^A. I. E. E. Transactions (1919), Vol. XXXVIII, Part 2, "Telephone Repeaters,"
by Bancroft Gherardi and Frank B. Jewett. . .
^A I E.E. Transactions (1923), Vol. XLII, "Telephone Transmission over Long
Cable Circuits," by A. B. Clark. Bell. Sys. Tech. Jour., Jan., 1923.
436 BELL SYSTEM TECHNICAL JOURNAL
line while speech is being transmitted in the opposite direction on the
other side. This device is known as an "echo suppressor." â– *
The enormous increases in long distance telephone traffic together
with the necessity of providing better transmission quality in con-
nection with radio broadcasting ^ and trans-oceanic messages, have led
to continuous design changes in telephone plant with more exacting
requirements for manufacture. To permit adequate and prede-
termined spacing of loading coils and repeater stations, the cable
design must be such as to insure definite capacitances per mile. There
must be a minimum of unbalance between circuits to insure that inter-
ference or 'crosstalk" is held to a low value. To handle the ever in-
creasing load of messages promptly and to secure further overall
economies, cables are being designed with a greatly increased number
of wire pairs, but of approximately the usual outside diameters to
permit the use of existing cable ducts. All of these design problems
are reflected in the machinery and methods of manufacture.
Manufacture of Cable ^
A typical long-distance telephone cable (toll cable) consists of
"quads" (double pairs) of paper-insulated electrolytic copper wire
(No. 16 to No. 22 B. & S. gauge) built up in layer construction and
covered with a lead-antimony alloy sheath 2f in. in diameter and | in.
thick. (Fig. 4.)
The raw materials for such cable consist of high-grade lead in pig
form, annealed electrolytic copper wire, and large jumbo rolls of
manila-rope wood-pulp paper. The first operation consists of slitting
the large rolls of paper into disk-shaped pads (Fig 5) . A sufficient num-
ber of these pads are placed in an insulating machine which applies the
paper to the copper wire in spiral form at a head speed of from 1,470 to
2,400 r.p.m. (Fig. 6). The insulated wires are paired very carefully and
then placed in a machine which first twists the pairs and then forms
them into twisted quads (Fig. 7).
The quads of wire thus built up are placed into a strander. One
quad serves as a center about which other quads are laid in alternate
layers as the material progresses through the machine (Fig. 8). Step
*A. I. E. E. Proceedings, Vol. XLIV, "Echo Suppressors for Long Telephone
Circuits," by A. B. Clark and R. C. Mathes.
^ Bell Sys. Tech. Jour., July 1930, "Long Distance Cable Circuit for Program
Transmission," By A. B. Clark and C. W. Green.
^ See paper " Recent Developments in the Process of Manufacturing Lead-Covered
Telephone Cable," by C. D. Hart, for historical treatment and developments
prior to 1927 — presented at the Regional Meeting of District No. 5 of the A. L E. E.,
Chicago, Illinois, November 28 to 30, 1927. Published in Bell Sys. Tech. Jour.,
April, 1928.
LEAD-COVERED PAPER-INSULATED TELEPHONE CABLE 437
by step it is thus built up, one layer being applied by each drum until
the full amount is obtained, after which an outer wrapping of paper is
applied to retain the insulated wires in shape and also serve as an
additional insulation from the lead sheath.
All telephone cable for local service (exchange cable) until recently
was made in much the same manner. Recently two new processes
have completely revolutionized its manufacture.
,. ^^
Fig. 4 — -Typical construction of long distance telephone cable.
Direct Application of Wood-Pulp Insulation
The process and machine recently developed to apply wood pulp
direct on wire combines the steps of paper making, slitting, (Fig. 5) and
insulating (Fig. 6) into one operation, and gives a continuous sleeve of
pulp paper around the wire.
Essentially, the process consists in forming simultaneously on a
438
BELL SYSTEM TECHNICAL JOURNAL
modified cylinder paper machine, 50 narrow continuous sheets of
paper, with a single strand of wire enclosed in each sheet, pressing the
excess moisture from the sheets, turning them down by means of a
rapidly rotating polishing device, so as to form a uniform cylindrical
coating of wet pulp around the wire, and then driving the water from
this coating by drying.
Fig. 5 — -Slitting of paper.
The material used in making this insulation is Kraft pulp, which is
prepared for use on the machine by beating as in the ordinary paper-
making operation (Fig. 9) and fed to the machine in a somewhat more
diluted form than in standard paper making practice.
In theory, the whole process is simple, but from a practical stand-
point, many interesting problems had to be solved before satisfactory
operation was possible. A continuous supply of wire must be fur-
nished, as it is not feasible to shut the machine down to change supply
spools. This was taken care of by removing the wire from the supply
spool by means of a flier without rotating the spool. This allows
time to braze the end of the wire from one spool to the next. Ordinary
annealed copper wire has a non-uniform surface due in part to the
residual drawing compound, h satisfactory surface is obtained by
LEAD-COVERED PAPER-INSULATED TELEPHONE CABLE 439
passing the wire through an alternating-current electrolytic cleaning
bath before it enters the paper forming machine. A narrow sheet is
formed on each conductor in an ordinary single-cylinder paper machine,
the mold of which has been divided into 50 parts by means of celluloid
strips and so arranged that a part of the sheet of paper is formed
before the wire comes in contact with it. The remainder of the sheet
is then laid down on top of the wire without any break in the formation,
Fig. 6 — ^Paper insulating machine.
and the resulting narrow ribbon of paper carries the wire imbedded in
it. Thus fifty conductors are being insulated simultaneously. Two
sets of press rolls take the excess moisture from the sheet, and leave
it ready for the polishing operation. Various types of polishers have
been developed and the one now in use consists of two short, specially
shaped blocks, with a third block located about centrally to the other
two. These polishers are rotated very rapidly around the wire
440
BELL SYSTEM TECHNICAL JOURNAL
(Fig. 10). Their construction is such that if an occasional lump or
break occurs in the sheet it does not cause clogging of the polisher.
Polished wet insulation carries about 70 per cent water by weight,
which has to be driven off by heat. The drier consists of a 25-ft.-long
electric box-type furnace, with heating elements extending the full
length of the top, and additional heating elements in the first 8-ft.
Fig. 7 — Twisting and quadding machine.
section of the bottom. These elements are thermostatically controlled
so that the temperature of the furnace can be set so as not to cause
charring of the insulation as it passes through the drier (Fig. 11). Two
spooling positions are furnished at the take-up for each wire, so that
as soon as one spool is full, the wire can be shifted to an empty spool,
and the full spool removed (Fig. 11). In this way, no shutdowns for
LEAD-COVERED PAPER-INSULATED TELEPHONE CABLE 441
Fig. 8 — Stranding machine.
Fig. 9 — -Beating equipment and pulp storage tanks.
442
BELL SYSTEM TECHNICAL JOURNAL
Fig. 10 — -Machine for polishing pulp insulation after its application to the wire.
Fig. 11 — Drying and take-up units.
LEAD-COVERED PAPER-INSULATED TELEPHONE CABLE 443
changing take-up spools are necessary. Individual wires are strung in
without shutting down. Tension devices are incorporated in the
take-up so as to avoid the possibility of any undue tension being put
on the finished wire. The normal speed of the machine is approxi-
mately 110 ft. per min., and the output per week is about 45 million
conductor feet.
The electrical properties of telephone exchange cables made from
this material compare favorably with those made from ribbon insula-
tion, and the annual saving per machine is an appreciable factor due
largely to the lower cost of raw material.
Improved Cable Stranding
Until recently all local cable (exchange cable) was built up or
stranded by the concentric layer method at a speed of 50 to 100 ft. per
min. (Fig. 8). This construction is being rapidly superseded by a unit
Fig. 12— 1818-pair unit cable.
444
BELL SYSTEM TECHNICAL JOURNAL
method, the first appHcation of which was made on the 1818-pair 26
B. & S. gauge cableJ
The unit method consists of two distinct steps. A flier strander is
used to strand pairs into individual color groups known as units, which
usually consist of 50, 51, or 101 pairs. A cabling machine then
assembles a definite number of these units into a round core form.
Thus the final cable size is some multiple of 50, 51, or 101 pairs. An
1818-pair cable built in this manner is shown in Fig. 12.
Fig. 13 — Flier strander.
The flier strander shown in Fig. 13 consists of a reel carriage or drum
for holding 101 supply reels of paired wire; a cotton serving head for
winding a cotton thread about the unit; a flier for stranding the unit;
a pulling mechanism or capstan for advancing the unit through the
machine; and a take-up for reeling the finished unit on a core truck.
By revolving the flier about the normally stationary supply it is
possible to obtain two twists in the unit per flier revolution. This
combined with the low inertia of the flier permits units to be stranded
at the rate of 300 ft. per min.
The cabling machine shown in Fig. 14 consists of 18 supply stands
equipped with suitable pneumatic brakes for holding and maintaining
tensions on the trucks of units, and a rotating capstan take-up. The
units are pulled through a distributer plate and covered with a pro-
tective wrap of paper. A twist is put in the cable between the dis-
'' Bell Telephone Quarterly, January 1929, " 1800-pair Cable Becomes a Bell
System Standard," by F. L. Rhodes.
LEAD-COVERED PAPER-INSVLATED TELEPHONE CABLE 445
tributer plate and the entrance point of the cable to the capstan. The
finished cable is taken up on reels capable of carrying three times as
much cable as the core trucks used with the concentric stranding
machine. These reels of cable are then handled through subsequent
manufacturing processes by electric trucks.
Fig. 14 — Cabling machine.
The principal advantages of this construction are that slightly less
copper and paper are required in large sizes of cable due to the shorter
lay in the outer strands. With the same investment in machinery and
building, a much larger production may be obtained. Much finer
gauges of wire may be stranded without danger of stretching beyond
its elastic limit.
Vacuum Drying
Dry paper is an excellent insulation for the conductors of a telephone
cable, but it must be bone dry. Dry paper takes up moisture rapidly
and 1000 lbs. loosely packed in a few hours will absorb 90 lbs. of
moisture in a room at summer temperature and 60 per cent relative
humidity.
A vacuum drying operation is applied to stranded cable prior
to the lead sheathing operation at a temperature of 270° F. for a period
of from 12 to 42 hours, depending on the size of cable. The vacuum
maintained toward the end of the drying cycle is less than 2 in. Hg.
The vacuum drying system installed at the Point Breeze plant has
446
BELL SYSTEM TECHNICAL JOURNAL
incorporated in its design many improvements in order to improve
cable quality, and also to reduce that part of the manufacturing cost.»
It consists of fifteen horizontal driers, each 40 ft. in length and 1\ ft. in
diameter, and one horizontal drier 40 ft. in length and 10 ft. 4 in. in
diameter (Fig. 15). The former driers are used for the ordinary toll
Fig. 15 — -Vacuum driers.
cable, while the latter single tank is used for drying submarine cables of
long lengths.
The drying ovens are arranged so that the loading end is located in
the cable room proper, and the unloading end in the dehumidified
cable storage room (Fig. 17). To prevent the exfiltration of dry air
from the storage room through joints between oven and brick wall a
novel type of seal is used. This consists of a flexible sheet of copper,
to allow for tank expansion, fastened and gasketed on the inner cir-
8 For further discussion and detailed factory layout of this system, see paper by
J. C. Hanley, Mech. Engg., March, 1931.
LEAD-COVERED PAPER-INSULATED TELEPHONE CABLE 447
cumference to the tank, and on the outer circumference to the brick
wall of the storage oven.
Auxiliary equipment used with the vacuum drying ovens consists
of two welded vacuum lines (twelve inches in diameter) vacuum
pumps, condensers and receiver tanks. A general view can be obtained
from Fig. 16,
Fig. 16 — ^Auxiliary equipment for vacuum driers.
One vacuum line is used to establish vacuum in a new tank load of
cables, and the second is used for maintaining vacuum in the tanks
once they have reached the proper point. The pump equipment
consists of four reciprocating feather valve vacuum pumps. The
pistons on these pumps have a diameter of twenty-nine inches and a
stroke of eighteen inches or a displacement of ten hundred and twenty
five C.F.M. The pumping capacity has been based on maintaining
absolute pressures of one-half to one inch in the vacuum tanks.
These values are based on a vacuum tank activity of eighty-five per
cent and on maximum leakage of approximately twenty pounds of air
into each tank through the door gaskets.
Two two hundred and twenty five C.F.M. surface condensers are
incorporated in the layout ahead of the pumps to condense moisture
given off by the insulated paper. Three thousand pounds of water
may be extracted in twenty-four hours.
New features incorporated in the oven are design changes of the
heater coil and tank. This coil, of which there are four in each oven,
consists of steam header inlet and outlet, instead of a continuous
448
BELL SYSTEM TECHNICAL JOURNAL
length of eleven hundred and twenty feet of pipe. This type of coil
not only makes a much neater appearance in the heating system due to
its rigidity, but also insures positive draining, with the elimination of
steam hammer, and also more uniform heating in all portions of the
tank. The tanks are completely welded instead of riveted. This
method of assembly insures a better average vacuum as well as elimi-
nating considerable maintenance work in caulking rivets, which
become loosened by the repeated expansion and contractions of the
drier.
Cable Storage Prior to Lead Covering
The air conditioned room (Fig. 17) is provided for the storage of cable
prior to lead sheathing in order to facilitate the covering of varying
Fig. 17 — Air conditioned cable storage room.
diameters of cable with a minimum of lead-press die-block changes,
and also to act as a reservoir for the fluctuating delivery of large
quantities of vacuum-treated cable. An alternative, that of storing
cable in the vacuum driers until ready for lead covering would require
an excessive investment in vacuum drying tanks and their operation.
The storage room, from which the cable is paid out directly to the
presses, is approximately 270 ft. long, 50 ft. wide and 12 ft. high,
and has been designed to prevent infiltration of moisture. Without
moisture proofing, the outside wet air would penetrate a concrete or
brick wall since the vapor pressure in the storage room is only approxi-
mately .007 in. Hg as compared to 1.02 inch outside the room on a hot
humid day. The moisture proofing was accomplished as follows: An
aluminum foil was placed over the inner surface of the outer portion of
the brick wall. This foil was suitably protected by a layer of saturated
rag felt and roofers asphalt. The remainder of the brick wall was
LEAD-COVERED PAPER-INSULATED TELEPHONE CABLE 449
placed in position over the moisture proofing membriine. The floor
was prepared in a similar manner.
The concrete ceiling of the room was covered with a layer of alumi-
num foil suitably overlapped and held in place by varnish.
As an added protection, all entrances are vestibuled and all cable
ports are equipped with air tight cable tubes leading to the presses.
When the press is not in use an air tight door is closed over the inner
end of the cable port.
Air Conditioning Equipment
The primary object in drying toll cable is to obtain as low con-
ductance and capacitance values and as high insulation resistance
as possible. This has a very important effect on the transmission
quality of the cable, and consequently justifies considerable expense.
l.t)
LlJ
^.
^
-^
(r
UJ
^
^
(ft
o
I
â– 5
.4
y
^
1
<r
/
y
2
/
INCREASE OF CONDUCTANCE WITH STORAGE TIME IN A
z
1.3
1.2
I.I
/
RELATIVE HUMIDITY OF 6 TO 7 PER CENT
LU
z
<
^
f
D
Q
Z
/
I-
20 40 60 80 100 120 140
HOURS OF STORAGE
Fig. 18 — Effect of moisture regain on conductance of vacuum dried cable.
A large amount of experimental work has been done to determine
the best methods of obtaining and retaining dry cables. At the end
of the vacuum drying cycle the cable paper is in such a dry condition
that its moisture regain when exposed to higher humidities is exceed-
ingly rapid. This is indicated by Fig. 18, showing the increase in
conductance over a period of hours when dry cable is exposed to
approximately 6-7 per cent relative humidity.
Working from these data and an estimate of the manufacturing ad-
450 BELL SYSTEM TECHNICAL JOURNAL
vantages from storage due to the elimination of lead press changes, it
was decided that a minimum moisture condition of .5 of one per cent
with storage periods not greater than 24 hours would result in minimum
conductance and capacitance values consistent with manufacturing
costs. The limit of .5 of one per cent was decided upon since to main-
tain humidities lower than that, costs would increase very rapidly and
entirely out of proportion to the change in relative humidity conditions
and the final result.
The air conditioning equipment installed at the Baltimore plant is
unique, in that a relative humidity of .5-.8 per cent is maintained at a
temperature of 100° F. without resorting to refrigeration. Silica gel,
highly porous form of silicon dioxide, or sand, is used as the water
absorptive medium. Before deciding upon this method of dehydration
other existing types of equipment were investigated. To obtain such
low humidities with the usual types of dehumidification systems would
require more than one stage of cooling and result in more expensive
operation costs in comparison with silica gel units.
The design requirements of this equipment were based on data
established for the following:
(1) Heat losses in the walls and infiltration of moisture.
(2) The movement into the storage room of core trucks filled with dry
cable at temperatures of approximately 260° F., and the incident
rush of storage room air into the vacuum driers when the
vacuum was broken.
(3) The loss of conditioned air when cables are being pulled through
the bell mouth openings to the press and also when the storage
room doors are opened.
(4) The actual moisture content of outside air, which must be dried to
replace losses in the storage room.
Based on a summary of the B.T.U. losses and gains which could be
expected in the manufacturing process, a study of the Baltimore tem-
perature conditions over a period of years, and an analysis of the
humidity conditions which would be encountered, equipment was de-
signed which will handle a volume of 13,000 cu. ft. of air per minute
amounting to a complete change of room air five times per hour. Of
this total amount approximately 10,300 cu. ft. is re-circulated, cooled,
dehydrated and brought back to the storage room requiring adsorber
capacity for only .6 pound of water per minute. Twenty-six hundred
cu. ft. of air is drawn from the outside to compensate for air losses at
various points in the room and to maintain an overall room pressure of
about \ ounce in excess of outside air pressures, requiring additional
adsorber capacity of approximately 4 jjounds of water per minute.
LRAD-COVERUn PAPER-INSULATED TELEPHONE CABLE 451
To maintain a normal operating temperature, it is necessary to
remove 17,500 B.T.U.'s per minute. This is accomplished by cooling
the air which is re-circulated plus the fresh air taken into the system
to 72° F.
The method of air distribution within the storage o\en was carefully
designed since the rate of regain of moisture by paper insulated cable
is dependent not only on the difference in vapor pressure of the cable
paper itself and that of the air passing over it but also on the velocity
of the air. The dry air is supplied through grill openings along the
side of the room at approximately 3-4 ft. from the floor, and at low
velocities consistent with positive circulation. Thus the driest air
is supplied at the point where it is most needed and, since the return
Fig. 19 — Silica gel drying unit.
ducts are located at the ceiling opposite to the grill openings, any
regain of moisture in the room itself is largely concentrated in air
strata above the cables.
Operation of the Baltimore conditioning system (Fig. 19) may be
described briefly as follows: Approximately 10,300 cu. ft. of air per
minute from the storage room is mixed with 2600 cu. ft. per minute of
outside fresh air. The temperature of this air mixture which may be as
high as 100° F. is lowered to a maximum of 68° F. by passing it over
and around copper tubes through which water at 58-60° F. is circu-
lating. The cool air then passes through the first silica gel adsorber
where it is partially dehydrated; then it is again cooled and is passed
into the second adsorber where the drying is completed and from which
452
BELL SYSTEM TECHNICAL JOURNAL
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LEAD-COVERED PAPER-IXSULATED TELEPHONE CABLE 453
it passes to the supply ducts in the storage room. The system is so
constructed that there are three simultaneous cycles (Fig. 20) : one in
which the silica gel is used as an absorbent, the second where it is
reactivated, and the third where a freshly reactivated bed is cooled to
68° F. Automatic controls switch the air currents into their respective
channels at established intervals. The condition of the vital parts of
the system is indicated continuously on a control board where tempera-
tures, air volumes, and relative humidities ^ are shown.
Lead Sheathing
The thoroughly dried cable core passes from the storage oven
through a tube, designed to minimize any exposure to outside air,
into the press where it receives its protective cover of lead. The
LEAD CYLINDER
\ %\ V4 V4 V^
Fig. 21 — Cross section of typical die block.
basic principle of applying lead sheath to cable is illustrated by Fig. 21
which shows a cross-section of a typical die block. This die block
consists of a core tube and a die, ring shaped, mounted in a hoUowed-
out block. This arrangement provides an opening adjacent to the
cable core which aids in definitely controlling the thickness and
diameter of the sheath. This die block is placed underneath a large
cylinder for receiving molten lead, and both are placed in a hydraulic
press.
In covering large cable, more than half of the total time is taken up
in filling the cylinder with lead and cooling it under pressure to a
point where it can be extruded. The tendency, therefore, has been to
build presses with larger lead containers, and in turn of larger capacity,
s Page 134, \'ol. 2, Industrial and Engineering Chemistry, April 15, 1930 — article
by A. C. Walker and E. J. Ernst, Jr.
454
BELL SYSTEM TECHNICAL JOURNAL
in order to make the productive time of extrusion a larger percentage of
the complete cycle of operation. Until recently presses were used
having a 30 in. diameter ram and a 42 in. stroke. Such a press has a
capacity of 1100 lbs. of lead per charge and extrudes a maximum of
4500 lbs. per hour. This type of press has the water ram located
below the floor line. The die block and lead cylinder therefore rise
slowly as the lead is forced out around the cable core. This varying
height of the cable as it is extruded in relation to the floor introduced
some difiiculties in the operation.
Fig. 22 — 34-in. inverted press.
The latest type of press used at Baltimore is illustrated by Fig. 22
and is known as the 34 in. inverted press. It was designed and
built by one of our outstanding American engineering firms. Its
stroke is 56 in.; the diameter of the ram is 10| in., with a lead capacity
of 1800 lb. per charge and a maximum extrusion rate of 5680 lb. per
hour. This press is approximately 21 ft. in height above the floor line,
and has the water cylinder mounted between the four columns at the
top of the press. The 34 in. diameter water ram has the steel lead ram
bolted to it. Connection is made from the water cylinder to a hy-
draulic pump, Fig. 23, supplying water at a maximum pressure of
LEAD-COVERED PAPER-INSULATED TELEPHONE CABLE 455
5500 lb. per sq. in. The four steel columns supporting these top
castings are 12^ in. in diameter. The steel ram e.xerts during extrusion
a pressure of approximately 59,000 lb. per scj. in. on the lead. At the
floor level of the press there is a cast-steel plate which carries a steel
spacing block upon which the die block rests. Above the die block is a
water-jacketed lead cylinder which is exactly centered over the feed
orifice of the die block. The die block and lead cylinder are held in place
on the cast-steel plate by four 2\ in. bolts. All these parts are station-
ary on this press, facilitating handling and inspection, and insuring that
the cable core always enters and leaves the die block at the same angle.
Fig. 23 — Lead press hydraulic pumps.
The concentricity of the sheath is affected not only by the contour
of the extrusion chamber, including core tube and die, but also by the
manner in which heat is applied; and the thickness is affected by
temperature and speed of extrusion so that the human element is an
important factor, and it is necessary to have thoroughly trained and
reliable operators on this kind of work. Temperature indicators are
used to show die-block temperatures, and the temperature of the
molten lead is automatically controlled and recorded.
Aside from increasing output, many studies have been made to
determine the exact mechanism of lead extrusion, the relative flow of
lead in dift'erent parts of the extrusion block, the effect of application
of heat at different points, etc.
As the lead-covered cable leaves the press, it is wound upon either
456
BELL SYSTEM TECHNICAL JOURNAL
wood or steel reels, depending upon its type. A full reel may weigh as
much as 10,000 lbs. These reels are rotated by means of power-driven
floor rolls which are controlled by the press operator's helper. After
the reel was tilled with cable, it was formerly the practice to push the
reels off the rolls manually. The latest type of floor rolls are equipped
with automatic ejector devices which lift one roll and cause the loaded
reel to roll oft' on the floor. This is done by means of a small hydraulic
cylinder connected to a pump which is operated by a valve mounted
adjacent to the floor rolls.
The Central Lead-Melting System
In order to supply the presses just described, large quantities of
lead-antimony alloy must be delivered frequently. The old and new
arrangements are shown in Fig. 24. With the old arrangement lead
loiQik loiQtoi ikOioi ikQioi loiQioi :o:Ooo: boo
CD acn nm u:2 a[:i] a[^ ac=i a
SPACE
FOR SKID
ADJUSTMENT
WITH INDIVIDUAL KETTLES (560 SQUARE FEET PER PRESS)
°o°°o° °o°oO° oOo°o; ;o;;o;
12 3 4 5 6 7 8
;o: :o:
9 10
II 12
cua c=]a cua c=ia czia u=ic^
SCALE
IN FEET
WITH CENTRAL MELTING SYSTEM (370 SQUARE FEET PER PRESS)
Fig. 24 — Space required for 34-inch inverted presses.
was delivered in skids by an overhead traveling crane to small melting
kettles adjacent to each pair of presses. This arrangement also
involved considerable manual handling, and introduced some variation
in the finished alloy sheath.
The new arrangement consists of melting all of the lead alloy in a
large furnace at a central location and distributing this molten lead
through a long-loop pipe line running back of the presses. Xear each
press a loop branch from this line is made and equipped with the
proper kind of control valve. This line is heated electrically and
the lead is in constant circulation. Such a system was built on a
small scale and tested under continuous operation for over a period of
si.x months, at the conclusion of which it was considered entirely
feasible to incorporate it as a part of our new Baltimore plant. In
order to take full advantage of such a system, the presses were placed
LEAD-COVERED PAPER-INSULATED TELEPHONE CABLE 457
close together, thus saving the space formerly occupied by the small
individual melting kettles, and the large central-supply kettles were
placed adjacent to the lead storage pit in order to minimize handling.
Views of this system now in use at Baltimore are shown in Figs. 25-28.
The details of this central lead-melting and distributing system will
be of interest to manufacturers using large quantities of lead or lead
alloy. Three oil-heated kettles are used (Fig. 25), and pipe and valve
arrangements have been set up so that the middle kettle is used for melt-
ing and preparing the alloy to the exact composition. The second kettle
Fig. 25 — General view of melting and supply kettles.
is used as a main supply and connected up to the distributing system.
The third kettle is a spare, and the piping is so arranged that it can be
used either as a melting or supply kettle. Each kettle has a capacity
of 120,000 lbs. of lead, and the melting capacity of the system is 80,000
lbs. per hour. Space is provided for a fourth kettle to take care of the
ultimate expansion of the cable plant.
Each kettle has two sets of low-pressure oil burners installed diag-
onally across from each other. An impeller type of vertical pump
having its intake about 12 in. above the bottom of the kettle, and
driven by a 20 hp. vertical motor, creates sufficient agitation by the
circulation of the metal to assure a uniform composition.
458 BELL SYSTEM TECHNICAL JOURNAL
The charging of the melting kettle with virgin lead is accomplished
by means of a specially designed lead-handling grapple (Fig. 26) which
has a capacity for 100 billets of the standard size or a total weight of
about 8500 lbs. Five to six of these charges or about 40,000 to 50,000
lbs. constitutes one melting cycle. The corresponding amount of
Fig. 26 — Charging of lead-melting kettle.
antimony is loaded into a special cradle which moves in a separate
chamber and is lowered below the surface of the lead, where the
antimony is dissolved by the washing action of the stream of lead
from the return line of the pump (Fig. 27).
The supply kettle is charged with the desired amount of molten lead
of the correct composition and temperature from the melting kettle by
means of the pump on the transfer line. Each kettle has one recording
LEAD-COVERED PAPER-INSULATED TELEPHONE CABLE 459
controller for regulating the temperature and one controller as a check
instrument and to actuate an alarm if the temperature goes above or
below a predetermined limit. Each instrument has its own thermo-
couple.
To reduce to a minimum the possibility of a prolonged shutdown
due to a breakdown in the lead conveying line, a duplicate pipe system
Fig. 27 — Antimony charging mechanism.
is provided which can be put into service in a short time in case of
failure of the line in use. The line ordinarily used is the one nearest to
the presses and is the service line while the line one foot to the rear but
at the same height, is called an emergency line.
The main-line piping system is made of seamless steel tubing sup-
ported on a roller-conveyor system to take care of the expansion and
contraction which amounts to 6| in. per 100 linear feet at 750° F. or a
total of approximately 20 in. under normal working conditions for the
460
BELL SYSTEM TECHNICAL JOURNAL
system. The down spouts are of seamless steel tubing and have a steel
valve at each joint with the main line and a service valve at one corner
of the "IT" bend. All joints are oxyacetylene welded, and no fittings
are used throughout the system. The lines are insulated with pipe
covering protected by a layer of fireproofed canvas (Fig. 28).
The lines are heated initially by a series of transformers which
supply a low-tension, high-amperage current directly into the pipe
by forming a loop of the supply and return line. Once circulation of
the lead has been established in the piping system, the main line
requires little additional heat from the transformers, as the flow of the
• -. '^^.•ci
Fig. 28 — Main lead supply lines.
lead will ordinarily keep the line up to temperature. Approximately
4 K\'A are required on each down spout while in use. The connections
leading from the transformer to the pipe are flexible, to allow for
expansion and contraction of the system.
Switches are provided on each building column opposite the presses
to enable an operator to shut down the pumping system in case of a
serious leak or failure of a valve.
This system has been in operation for about nine months and has
resulted in a higher quality of lead sheath due to more uniform compo-
sition maintained. In addition there are considerable savings in fuel,
reduction in dross, and elimination of a large amount of heavy manual
effort. The press room is now clean and cool, resulting in much better
working conditions and in turn an indirect improvement in the
quality of the product.
LEAD-COVERED PAPER-INSULATED TELEPHONE CABLE 461
Testing Lead Covered Cable
After the cable is stranded each conductor is tested from end to end
for continuity and against every other conductor for crosses. Defects
are repaired and after the cable core has been dried the lead sheath is
applied. After the application of the sheath the cable is allowed to
stand until it cools to room temperature. Fig. 29 shows the cooling
floor and test mezzanine in the Point Breeze cable plant. The reels of
cable issue from the lead presses at the right; are cooled in the central
area and tested beneath the mezzanine at the left.
Fig. 29 — ^CooIing floor and test mezzanine.
When the cables are cooled the conductors are given a final test for
opens and crosses which may have developed due to strains imposed
during the sheathing process. Most toll cables have a number of spare
wires and if fewer than the allowable number of above defects are found
the cable is tested for dielectric strength, insulation resistance, mutual
capacitance, capacitance unbalance and defects in the sheath. Die-
lectric strength tests are made between each conductor and every other
adjacent conductor to which failure may occur and between all
conductors and the lead sheath. The potential used for these tests
ranges between 350 volts, A.C., the lowest value used for certain con-
ductor to conductor tests and as high as 5,000 volts, A.C. for some
conductor to sheath tests. In making the conductor to conductor
tests a large number of circuits are involved so that interesting prob-
462
BELL SYSTEM TECHNICAL JOURNAL
lems arise in designing switching devices to apply the test potential
between all conductors.
Defects found by continuity, cross or dielectric strength tests must
be located within the cable in order that repairs may be made. The
point of break in open conductors is located by comparing the capaci-
tance between the defective conductor and the adjacent conductors
with the capacitance between a conductor known to be good and its
adjacent conductors. Preliminary locations of crosses between con-
ductors and between conductors and the sheath are made by means of
the modified Murray Loop test. Final locations are made by means of
a search coil and telephone receiver which responds to currents of
audible frequency circulated through the crossed conductors.
Fig. 30 — Typical test set installation at Baltimore plant.
Fig. 30 shows a closer view of a section of the test mezzanine at the
left of the cooling area. The test desk in the foreground is designed
for making insulation resistance, D.C. capacitance, A.C. capacitance,
and conductor resistance tests. The test desk in the center is a
shielded precision bridge for making capacitance and conductance
measurements at audio frequencies. Two test desks in the back-
ground are capacitance unbalance bridges. All desks on the mezzanine
floor are provided with test leads which terminate in outlet boxes on
the test floor below.
Figs. 31 and 32 show a front and rear view respectively of the
LRAD-COVRRRD PAPER-TNSULATKD TRLFPHONE CABLK 463
insulation resistance, D.C. capacitance, A. C. capacitance meter, and
conductor resistance test desk which appears in the foreground of
Fig. 30. Insulation resistance measurements are made between
conductors and between all conductors and the sheath by observing
the deflection obtained with a high sensitivity reflecting type D'Arson-
val galvanometer through which a potential of 500 volts D.C, is
impressed on the insulation of the conductors under test. Due to the
Fig. 31 — D-C. insulation resistance test desk — front view.
high insulation resistances involved and the extreme sensitivity of the
measuring circuit, considerable difficulty is likely to be encountered
with leakage in the test apparatus itself, especially during times of high
relative humidity. To overcome this source of error special test
circuits have been designed which employ a shield (Fig. 33) to eliminate
from the measurement all extraneous leakage other than that of the
cable. The direct reading capacitance meter is used extensively for
464 BELL SYSTEM TECHNICAL JOURNAL
mutual capacitance measurements where the highest accuracy is not
essential and where conductance readings are not desired. D.C.
capacitance tests are made by the charge and discharge method,
employing a ballistic galvanometer. In general, D.C. capacitance
tests are not fully indicative of the characteristics of the cable at
telephonic frequencies and for this reason are not extensively employed.
Fig. 32 — D-C. insulation resistance test desk — back view.
Conductor resistance tests, Fig. 34, are made by means of a Wheatstone
bridge circuit specially arranged to read directly the conductor re-
sistance per mile at 68° F.
Although the majority of mutual capacitance measurements are
made by means of the direct reading capacitance meter, the capacitance
and conductance of a percentage of all cables are measured at a
frequency of 900 cycles per second by means of the shielded capacitance
LEAD-COVKRED PAPER-INSULATED TELEPHONE CABLE 465
bridge.'" Due to the fact that these bridges are frequently employed
in shop areas where some noise exists it has been necessary to develop a
GALVANOMETER
AYRTON SHUNT
rM-
^SHIELD
ALL OTHER
CONDUCTORS'
CONDUCTOR
UNDER TEST
<0.l MEGOHM ^=="
> PROTECTIVE
<" RESISTANCE
-=- 500
^ VOLTS
-[-DC ^
Fig. io — Shielded insulation resistance test circuit.
device to replace the telephone receiver as a means of indicating
bridge balances. The visual bridge balance indicator used consists
essentially of a vacuum tube circuit in which the alternating current
— NOTES —
1 - SET RESISTANCE OF (A) EQUAL TO LENGTH OF CABLE
2 - BALANCE OUT LEAD RESISTANCE WITH (B)
3- READ OHMS PER MILE AT eS^F FROM (C)
SWITCH FOR MAKING
PRELIMINARY BALANCE
Fig. 34 — Conductor resistance measuring circuit.
" 5e// Sys. Tech. Jour., July, 1922: "Measurement of Direct Capacities," G. A.
Campbell. Transactions A. I. E. E., \'ol. XL\'l, May, 1927: "High Frequency
Measurement of Communication Apparatus," W. J. Shackelton and J. G. Ferguson.
466
BELL SYSTEM TECHNICAL JOURNAL
MILLIAMMETER
Fig. 35 — -Visual bridge balance indicator circuit.
input to the indicator is amplified and the rectified output is indicated
by the reading of a D.C. milHammeter. When the bridge is balanced
there is no input to the indicator so that the milliammeter pointer
Cb+^bm
GROUND l||| 1 1 ( B
cb
PHANTOM TO WHITE SIDE UNBALANCE:
AGROUND 2 [C| + C2-(C3+C4)+2-(Cw-Cwm)j
PHANTOM TO BLACK SIDE UNBALANCE^
2[C| + C4-(C2 + C3J + ^(Cb-Cbm)]
BM) i| |||i GROUND
Cbm
•CwM
^^ GROUND
Fig. 36 — Phantoni-to-side capacitance unbalances.
LEAD-COVERED PAPER-INSULATED TELEPHONE CABLE 467
returns to the lower end of the scale, See Fig. 35. The high resistance
in series with the grid of the third or rectifier tube prevents the over-
loading of the milliammeter when a large input voltage is impressed on
the indicator circuit.
Toll cable in addition to the above receives a capacitance unbalance
CWM'
C4
:cb
Cbm
C3
C2
Ci
GROUND
AGROUND UNBALANCE = (C2 + C4) - (c | + C3)
I ||ll- GROUND
cbm
'GROUND
Fig. 37 — Side-to-side capacitance unbalances.
test which is indicative of the cross-talk existing between circuits.
These tests are made with a special shielded bridge mentioned above,
which measures the capacitance unbalance between side and phantom
and side circuits of "quads" (Figs. 36 and 37). These bridges are also
provided with visual balance indicators as described above.
After the cable has successfully met all electrical requirements both
468
BELL SYSTEM TECHNICAL JOURNAL
ends are sealed and the cable is prepared for shipment. Certain
types of cable receive an additional gas pressure test to detect minor
defects in the lead sheath which otherwise may have escaped attention.
Dry nitrogen is forced into the cable to a predetermined pressure and
the cable is allowed to stand for a specified period. Loss of pressure
during the test period indicates that the sheath or seals contain one or
more defects.
Armoring of Telephone Cable
Two types of armored telephone cal)le are in use, Fig. 38. Subma-
rine telephone cables for ri\'ers and harbors are usually protected by
TAPE ARMORED CABLE
SUBMARINE CABLE
Fig. 38 — Typical construction of tape and wire armored cables.
layers of jute and wire placed on the outside of the lead sheath.
This type of armor is quite familiar and is called wire armoring.
Cable buried in a dirt trench is armored in a similar way except the
wire is replaced by two layers of steel tape. This is called tape
armoring. It is adapted to certain localities where there are long
stretches of open country and the conditions indicate one or two cables
will handle the requirements for a considerable number of years.
A typical wire armor is made up of a bedding of 100 or 150 pound
jute roving, impregnated with suitable preservative after serving, by
passage through immersion troughs, over which a layer of armor wires
is applied. In some cases, a covering of outer jute flooded with coal tar
is used. When an unusual degree of protection is desired, a second
layer of armor wire is applied. In such cases a bedding of jute is used
between the layers.
Recent trends in the design of wire armored cables are leading
toward cables of nmch larger diameter. At the Point Breeze plant
there is an unusually large wire armoring machine (Fig. 39). It is
designed to handle cable up to 5% inches in diameter over the armor.
LEAD-COVERED PAPER-IXSULATED TELEPHONE CABLE 469
Tape armored cable differs somewhat in construction depending; upon
the kind and diameter of cable armored. A typical design is made up
as follows: A coating of asphalt is first applied to the cable and over
this a layer of impregnated kraft paper. Another layer of asphalt
compound is put on and then two servings of impregnated jute roving
^
]
V *li
H^H
i^HH
'^'i^^fl
X
to
N3i
m
3
^J^BJBE^^'"* ^"^
i^^lHIl^ .
Jf^
»~--ft.
w
^^^^
aSJl» ^-^.^^ -J
^^3^
^^
^
raW
^pr __
â– ^
f\ A
^^^AiflH SMBHI
mM
jljjk -JtiV
Hk
Mij^
^gi. ^IS^^^gfSn
W~^'
^I^H Iff^^n ^fSH^^S
wBBeKp^
j^^
W^i
i
wm
H
â–
Fig. 39 — ^Wire armoring machine.
with opposite directions of lay. Asphalt coatings are used between
the two servings and on the outside of the second. Next two steel
tapes are served with the same direction of lay and with the second
tape overlapping the gap between the edges of the first. Again the
cable is given a coat of asphalt. One serving of impregnated jute
roving, a coating of asphalt and a layer of impregnated jute yarn
with opposite direction of lay are next applied. An application of
non-adhesive compound composed of whiting, glue, and water com-
pletes the armor coating. The machines used for tape armoring are
shown by Fig. 40. They consist of a supply position for the lead
sheathed cable, asphalt tanks, paper heads, jute heads, two steel
tape heads, a capstan and take-up. Tanks for melting the asphalt
compounds before their use in the machine are also provided. This
470
BKLL SYSTEM TECHNICAL JOURNAL
type of cable is protected from mechanical injury and soil corrosion, and
can be laid very quickly and cheaply. One interesting advantage
gained through the use of this type of armor is that a magnetic
shield is thus placed around the cable greatly reducing the efTects of
induction.
£'J.rmf^^.'^::kM
Fig. 40 — -Tape armoring machine.
Conclusion
The application of scientific and engineering elTort to improvements
in the processes and machine equipment for manufacture of telephone
cable is fully justified by the results which have been obtained from
both an economic and quality standpoint. New raw materials and
alloys together with new designs of cable will be forthcoming in the
future in the effort to improve and extend the long distance telephone
service. New communication devices will be invented and perfected
for use in connection with such cable and these in turn will have a
radical effect upon the cable design, the process and the equipment for
its manufacture. I'he engineers and scientists engaged in such
manufacturing activities are indeed rendering a broad service not only
to the men and women employed in the immediate industry but also
to the people at large who use these facilities.
LEAD-COVERED PAPER-INSULATED TELEPHONE CABLE 471
In concluding, the writer wishes to acknowledge the efforts of the
men who have carried these developments to a conclusion, in particular
Mr. H. G. Walker on the pulp wire process; Mr. L. O. Reichelt and
Mr. H. J. Boe on the unit cable machinery; Mr. H. F. Carter on the
central lead melting system; and Mr. J. Wells on the air condition-
ing system.
o
Effect of Ground Permeability on Ground Return Circuits
By W. HOWARD WISE
The formulas for the self and mutual impedances of ground return
circuits are derived without restricting the ground permeability. Curves are
given to show the effect of a ground permeability 1.7 on the mutual imped-
ance between two parallel ground return circuits with the wires lying on the
ground.
X account of the irregular and heterogeneous character of the
major portion of the earth's surface and the consequent difficulty
in choosing a conductivity to be used in a computation of ground
return circuit impedance it has heretofore been considered useless to
take into consideration the possibility of an earth permeability greater
than unity. However, since the permeability may sometimes be
known to be appreciably different from unity and it is always desirable
to reduce the probable error in a computation and since the inclusion
of the permeability in the formulas may sometimes lead to a better
agreement between the theory and experiments it seems worth while
to provide formulas which include the permeability.
The self impedance of a ground return circuit is
Z = r: + iico log p":a + MP + iQ),
where z + /2co log p"!a is the self impedance with a perfectly con-
ducting ground and 4w(P -f iQ) contains the effect of the finite con-
ductivity and permeability of the ground. Carson * has derived an
infinite integral and series expansions for P + iQ on the basis of unit
permeability. The infinite integral derived here is arrived at merely
by going through Carson's paper and writing in the permeability
wherever Carson has replaced it by unity. The reader will be expected
to have a copy of Carson's paper at hand as not all of the steps in his
paper will be here reproduced.
Equations (23) and (24) respectively are the new infinite integral
formulas for self and mutual impedance. Equations (A) and (C)
respectively are the new asymptotic and convergent series formulas
for P and Q. The functions m and / occurring in equations (C) are
functions of the permeability. Since some of them are defined by
series and their computation is consequently rather laborious, enough
1 John R. Carson, Bell System Technical Journal, Oct., 1926.
472
EFFECT OF GROUND PERMEABILITY 473
of them are tabulated for values of (x from 1 to 1.7 to provide for the
computation of P and Q for values of ry up to 2.
Equation (1)^ is unchanged but there is a new definition for a
a = 47rX/xco.
Since curl E = — {dldt)yLH equations (2) and (3) have the factor ^
added to their left hand sides.
The next change is in the application of the boundary conditions.
At the surface of the ground H,- and filLj must be continuous. The
equations to be solved for F{t) and 0(r) now become
1
whence
-^ Vr'' + la F{t) = 2/e-''- + 0(r),
^tF{t) = lie-''-' - cP(t),
^w-,/;-^': 47, (11)
'Vr'' -\- la -\- IJ.T
4>{r)= '^;]^'^- ^e~"r2L (12)
Vr- + la + jXT
The new equations (13), (14), (18), (19), (20), (23) and (24) are
£, = - 7'4co/m r ^"^.•^ " - f-r''+W^^+^^^^ (13)
Jq vt' + ia + /UT"
(14)
• 00 ,,-(ft'+2/')r
Ez — — lAo^Ifx I , cos .v'rJr
f2c./log^-^F, (18)
p OS
'00 p-{h'+p')T
^•00 ^-(ft'+2^')T "
'/4w/m . ^r - i2a;/log— + TV, (19)
Jo Vr" + i + ;ur ^
n = (G + tcoC) r z + /2co log^'
-]- 7'4a;/i
-\- I + tXT
, (20)
;74 BELL SYSTEM TECILXICAL JOURNAL
K + iX = Z = s + 72co lop: h -^â– 4coM , di
« Jo \^^^^T^ + Mr
//
= s 4- /■2co ]o^ — + 4a)(P + iQ),
Z,o = 72a) loc; -7 + /-Ico/i I , cos .t't(/
(23)
(24)
= i2u>\og^ + 4a;(P + iQ).
P
The principal steps in the derivation of equation (18) are given in
Appendix I.
The new definition cf P + iQ is
P + iQ = //i I -^=^ COS .v'rJr.
Jo V?" -h / + /ur
Replacing / by i"^ and assuming that v is a real quantity this is
/^QO -p(/i'+:/'+ix')r
P + iQ = fxz'-R â– dr,
Jo V'' ^ 1 + I^T
where R is used to indicate that the real part is to be taken.
The asymptotic expansion is easiest derived by expanding
1/(Vt^ + 1 + fir) into an ascending power series in t and integrating
termwise.
l/(Vr-^ + 1 + Mr) = 1 - MT + (jJ.' - |)r^ - (m' - m)^'
whence, writing /;' + y' + '-v' = n^ ,
P + iC) = txl'
COS COS 2B , cos 30 ,
M — 7^; — r (m- — 2) — —r ^ •
cos 50
(m^ - m)^3! + (m^ - ^^ + I) ^^4!
, ., ...cos 69 .
whence, separating the real and imaginary parts,
P =
^V r
cos I /^ , .V COS 30
+ (2m" — 1) - — ^ —
+ 3(12m'-'-8m^-3)^^ +
EFFECT OF GROUM) PERMEABILITY
475
At"
1 ! cos 26
+
1 Y
5! cos 60
9! cos 100- +
Q =
M L
cos Q .- , .. cos 30
(^)
+ 3(12^2 - 8m^ - 3)
cos 50
x2 - 1 \2
3 ! cos 40
- W
I ! cos
+ ( ^^^— r^ V 1 1 ! cos 120 - +
It is worth noticing that when /- is so large that only the leading
terms in P are of importance
P = [m + (//i + //,..) V27rXaJAt]47rXco[.v- + (V/i + //,.)-;
At power frequencies (Jii + Ji2)\2wXo:iJ. is small in comparison with fx.
When /i — 1 is small a series in powers of /x — 1 is a convenient
torm of solution. This is readih- arrived at bv writing
1 + MT
I -I 7 L
\>T 1 + r/
1 ( "
V
â– -h â– â– â–
' \ VV ^ , -
h J
The expansion is absolutely convergent for all values of r if
e = M - 1 < 2.
â– )^
l.XVr^ + 1 + M^) = (Vr- + 1 - r) - er(\r- + 1
+ eV2(Vr-^ + 1 - r)-^ - +
e3(8r'
= \72 + 1[1 + e272 + 62(4r^ + t2) + e3(8r6 + 4r^)
+ e^(16r8 + 12t6 + r^) + ^^ilr^^ + 327^ + 676)
+ t«(64ri2 + 8O7IO + 2478 + 76)
+ e7(1287" + 1927^2 ^ 807'o + 878) + • • •]
- [r + €(27^ + 7) + e-(47^ + Zt'^)
+ 6^(877 + 8t5 + 7^) + 6^1679 + 207^ + 57S)
+ 6^(32711 + 487^ + 18t7 + 7^) + • • •]•
476 BF.LL SYSTEM TECHNICAL JOURNAL
Writing c = v{h' + y' + '-v') = vrt^^ we have then
P + 'Q = >"'-'< I v?TT + ./ -
(5)
J-»CO
K^(c)
(â– ^ S i''5 "^ .^2527
cosh- </) r-^s>"h
- + •••;
jui'-i?[/(r) — (l/c^)] = M times Carson's P + i(2 with a = /j-AirXw.
The problem is now reduced to the tedious procedure of differen-
tiating J'{c) and separating real and imaginary parts twice for each
power of e. The chief steps are given in Appendix II. The result
is best written in the form
P =
Wo
W4 / r.y 1^ , "'
- ( - 1 cos 80
â– mn { !'\
+
2 Ll!2!V 2/
6!7! V 2
cos 120 +
''''^'^-jwA^J "'"^'^
V2
Wi
ri cos d
3
+ 5!6!ilj smlO0- +
r-[^ cos vS(9 ^1^ cos 50
r/ cos 70 _
+ W7 3252729 +
^1^ cos 60
, ri2 cos 20 / , , , 2
/fi + w/glog
, ri'OcoslO0/. , ,2
+ 2^^516! (^^o + w„log-)- +
(O
Q- -
nt'.
l!2!Vl^-^'^ln 1'"^""
+ ^ 1' -««- +
2 See Jahnke & Emde, "Funktionentafeln," pages 171 and 93.
EFFECT OF GROUND PERMEABILITY
477
sin Ad
sin 80
+
vui rx
6!7
-^ ) sin na
+
+
<iV
Yx COS B , ^i''' COR 3/? ^i^ COS 50
Wi :; h "';i T^T^ " — ^'5
325
?;/7
3-5.27
r^ COS 70 + +
3252729
, 1 // , 1 2\ r,4cos40/, . , 2
ri^ cos 80 / 2 \
+ -i^i!5r(v^^ + '"^'^*^^ 7;; - + •••'
where ri = ^/Vm = V47rXa)|_.T2 _|_ (^/^ _|_ yj2] = Carson's r and the per-
meabihty is contained in the functions ni:, and l^.
The definitions of the nu and /r will be found in Appendix II. The
table of numerical values should suffice for most needs.
TA?iLE 1
M
-\u
h
h
u
h
/lO
1
0.03861
0.67278
1.08945
1.38112
1.60612
1.78945
1.1
0.04619
0.70382
1.23834
1.71141
2.1758
2.6549
1.2
0.05264
0.72954
1.38429
2.07062
2.8568
3.7890
1.3
0.05808
0.75059
1.52655
2.45663
3.6558
5.2371
1.4
0.06261
0.76756
1.66456
2.86745
4.5785
7.0466
1.5
0.06631
0.78095
1.79799
3.30122
5.6305
9.2669
1.6
0.06923
0.79121
1.92660
3.75623
6.8167
11.9492
1.7
0.07159
0.79871
2.05026
4.23089
8.1417
15.1467
M
«o
MZl
m2
mz
W4
ms
1
1
1
1
1
1
1
1.1
1.04762
1.06700
1.09751
1.13529
1.17851
1.22625
1.2
1.09091
1.12837
1.19008
1.26928
1.36318
1.47078
1.3
1.13043
1.18469
1.27788
1.40147
1.55291
1.73236
1.4
1.16667
1.23643
1.36111
1.53153
1.74676
2.00996
1.5
1.20000
1.28403
1.44000
1.65922
1.94400
2.30254
1.6
1.23077
1.32793
1.51479
1.78442
2.14402
2.60929
1.7
1.25926
1.36845
1.58573
1.90706
2.34630
2.92942
M
We
nn
m%
m9
W7iO
Wn
1
1
1
1
1
1
1
1.1
1.27799
1.3334
1.3926
1.4554
1.5217
1.5918
1.2
1.59192
1.7270
1.8767
2.0420
2.2241
2.4244
1.3
1.94162
2.1834
2.4613
2.7798
3.1441
3.5602
1.4
2.32690
2.7055
3.1556
3.6895
4.3217
5.0696
1.5
2.74752
3.2957
3.9685
4.7925
5.8003
7.0324
1.6
3.20326
3.9566
4.9089
6.1108
7.6264
9.5370
1.7
3.69388
4.6903
5.9856
7.6674
9.8497
12.6816
BELL SYSTEM TECHNICAL JOURNAL
m
EFFECT OF GROUND PERMEABILITY 479
The curves show the effect of m = 1-7 on the mutual impedance
between two parrallel ground return circuits with the wires lying on
the ground. The dashed portions of the curves were not computed.
Appendix I
Equations (4), (7) and (17) substituted into (16) give
E.Cv, y) = £,(.v, 0) - '"j[''[ ,4V-y)'
/•«
+ (/)(r) cos XT-e-^\l
Jo
= E,{x, 0) + /CO i 4>{t) cos .Tr(f-^' - 1) —
Jo ^
+ 7 CO 7 log
7 ^^
T
x^ + (h - yY _ dV
X^ + //2 dz
+ M7"
..-, C^ M^" COS XT J
Jo Vt- + ia
Jo -yjT^ -f ia + yur ''"
x2 + (//.- 3^)2 aF
+ icci log
pte
x2 + h^
Jo V^
-(.ft + V)T
Ho:! I , COS xtcIt
+ 7a -|- /XT
J""" . /h , N h X COS .rr ,
. ^, .1-2 + (// - yY dV
+ 7co/ log ^ — ; — 7T ^
^ x- 4- //2 dz
= — 74co/ I , COS XTCIT
Jo VP
+ ta -\- yur
X-2+ (// - yY dV
+ 7coi log -^—j — 77 — ^ 75 — ^— •
x^ + (// + v)^ dz
Appendix II
The succeeding analysis has been considerably shortened by writing
In — I
480
where
/(O = -, +
BELL SYSTEM TECHNICAL JOURNAL
n J
1 S
COO
r r
+ \
+
3 325 32527 3252729
1 /r\2
+ -
.Cio - i20YJ2l I 2 y + ^'•■"'213! V 2
1 / cV
S40
3!4! V 2
+ -
1!2!\2/ ^2!3!V2
/(")(,) =
1 /£V+ ...
3!4!\2/ ^
_ '"8 7^
(;/ + 1)! 1 (" - 1)
c"+2 ' 2 c"
1 (;; - 3) ! 1-3 (n - 5) !
2-4 c"-- ' 2 •4-6 c"-^
1-3-5 ••• (n - 3) 1!
"^ 2 •4-6 • • • « t2
+ (-2)"/2
-
(w/2)!r (l-\-n/2)\c'
3-5
• 7 • • • (w + 3) 1 1325 • 7 • • • (w + 5)
, (2+w/2)!^5 -1
"^2 132527 -9 •••(« + 7) "^
/ l\("/2)+l
1-3-5 ••• (;? - 1)
~ "^)
0!(^ + l)!
\ - /
~ fl(n/2)
3-5-7 ••• (;/ + 1) /c\2 ,
/ J \ («/2)fl
"1-3-5 •••(;;-!) 3-5- 7 •••(« + 1)
I 2J
0!('i+l)! ,!(i; + 2)!
-«)'
+ ^^
7-9 ••• (?; + 3) / rV
2-(^ + .)! ^^^
log —
7(
the 71 being an even integer.
The inverse powers of c all cancel out, in equation (B), and there
EFFECT OF GROUND PERMEABILITY
481
remains
P + /() = ixv-R \-jh - ^^ g.i + 3^ gs - 32^2729 97 +
^\
325
"^ '" oTT! ^° ~ ^ -° TI2T 2!3! ^^
- fi
(cjlY
3!4!
/'6 +
+ ^ log —
9o
(r/2)2 , (r/2)^
1!2!
§2 +
213!
94
3!4!
+
(^^)
]}•
where
iiopo — S 10 — f^i2i J 2\ "^ ^^ \ '^ ^- 2^ • V ~ ^"^
,, ^^ 1-3-5 ,^ 1-3
+ e^ 16r
23-4!
1-3-5-7
2^-5!
3
2-- 3!
12r
^20 777^2 = faOTi - «2i-3i--^ + €- 4^'
2!
2-3!
1 • 3 • 5
*3 23.4!
3-5
2-2!
+ r:
1-3
22-3!
- +
i31
2--4! '''2-3!
3 • 5 • 7 _ , 3-5
2«5! 'ii'i^ 22.41
1 1 o. 5 , ,/_ 5-7 . 5 \
r30 3|^4 - rsojj - «--^4i2T^+ '" \ -^^^22^! ~ -^^'I^/
— e O.U3
5-7-9
^-4f,2^
5-7
5!
+
+
9o = 1
-tT+^^t^
1-3 1
22 • 3 ! 2 • 2 !
i:^-4-i:A,+
23-4! 2'^-3! ' ^
1 -1- 9-A_ : 2/4l:l__i_
2!^' " 2! 2-3!"^ ' I 2-'-4! 2-3!
^3-5-7 , 3-5 . ,
482 BELL SYSTEM TECHNICAL JOURNAL
1 1 , ^ A. â– >(a ^â– '^ ^
3!^* " 3! 2-4! ' V 2-'-5! 2-4!
21.1' / 92.71 71.]
>■)• / :)
93.21 92.91
'•^' 8^^- 4V^ ) +
9' -2' / 2--3' 91-91
l!g., = 1! - 62^=^+ 6-4 • -
7 \ 7-9 7
93.41 22-3'
7-9-11 7-9
2' -3! / 2- -4! 2' -3!
2\q:, = 2\- e2-— +.M 4
9 V 9-11 9
2^-5! _42!Jd,+ _
9-11-13 9-11
The way in which the succeeding terms of each series are to be
formed will be made clear by comparing the numbers just preced-
ing the fs in the p series with the numbers in the expansion of
1/(Vt' + 1 + ixt) into a power series in e.
The series converge if e = ^u — 1 < 2.
The q series are all represented by the single formula
/ 2 \i+(^/2) ^/x , X ^ . X e
1 + M/ \2' 2' " ' 2' 2
2 \l+(x/2)
1 +M
1+ ^ /^^--2
.V + 4 V 2 / 2
xix + 2) / e\-(.v - 2)(.r - 4)
+
{x + 4){x^b)\2j 2-4
xix 4- 2) / e\=^ (.V - 2) (a- - 4)(.v - 6)
(.r + ())(x + 8) V2/ 2-4-6
x{x + 2) / 6 y (a- - 2)(.v - 4)(.v - 6)(.v - 8)
■^ (A- + 8nA- + 10) V2/ 2 •4-6 -8 "^
I
good for all values of e if a- is even ; good for e :^ 2 if a- is odd. Other
series are available for odd x and 2 < e but there is little likelihood of
their being needed.
EFFECT OF GROUND PERMEABILITY
The p series are all comprised in the single formula
"~ e2f(2+x/2)l-
483
/ , , x\ ,{ ^ 1
^O+x/2)0px — I 1 + :^ I ! ! 1(1+1/2)0 -,
V 2; I ^^^
.r+1
2/ '
2( 2+^H
12/1. (.r+l)(x + 3) x+l
~r ^ I •*<; (3+1/2)2 -/ ;;7\ i(2+i/?>i
2M 3 + ^ 1 '
2( 2+^»'
Since '{„m = r«(n-i) + wz^n-i-m) wc Can write this
i(l+x/2)0px — Wi(Zi + 5i,
where
^x — ( 1 + ^ ) M r( 1+1/2)1/2
+ e2/4r
1
i+i"
(3+1/2) (2+1/2)
— e2f(2+x/2) (1+1/2)
(.x+ l)(x + 3)
2^(^3+1)!
.T + 1
2( 2+^H
— f(2+x/2) (1+1/2)
X + 1
2( 2 +1 i '
5o =
M 1 1 + M
log— ;S —
ix^- 1
2
/ - r/1 , 1 2m ,_ 1+)u
^^^ irr^j (2 + ^:^-07^^^^°^^-
1 :x; 4- 2
^x = -. 7 7 (fx/2(x/2-l) - 5x-2) for 4 < X.
]r- — \ X — \
Bv separating the real and imaginary parts one gets from equation
'^'~'^\[~iS '''''' ^^''^'^i^\{h ^ co^^^-5«- +
+ 2M
M_
V2
^ (-')"sin 20-52 --|^ (I Ysin 60-36 +
r cos
31
H COS 30 y^ cos 50 , ,
93 — :^^^^- 35 + +
^^â– 5
2
+ ^Tl^(lj^^«^20(r2oi>2 + g2log^^,
1 M / ''\^ / 2
~ 2 3!4! i^ 2 j ''''" ^^ V ^''°^' "^ "^^ ^"^ T^
+ \ 5!^! (l)'"""' 100 (uoPio + gio log I;;) - + •••.
4 8-4
BELL SYSTEM TECHNICAL JOURNAL
= -i.
+
V2
yj^^^ j ™s2^g2 -^(^j COS 6^56+ - ••• J
-^, ( 0'sin 40-54 - 4^ ( 2'y-i" «^-^s +-•••]
r cos 6 , r^ cos 36 r^ cos 56 , , 1
— r-^^^ + ^^5-^^^' - ":;^5^^^^ - + + •••]
~ 2 2!l! ( ^"'' "^^ ( ^''^' + ^' '""^ Vr )
Equations (C) are now got by writing r = riVM, w?^ = ^u^+^-^'^^g^ and
4 = Wz(cOx - log. tVm) + m'+^^/^'S:,
log« 7 = 0.5772157.
Negative Impedances and the Twin 21-Type Repeater
By GEORGE CRISSON
This paper discusses negative resistances and impedances. It describes
their properties and some devices by which they may be produced physically.
Certain properties of negative impedances when used as series and shunt
boosters for amplifying speech waves in telephone circuits are discussed.
The paper concludes with a description of the circuit and properties of the
twin 21 -type repeater.
WHEN an e.m.f. is applied to the terminals of an ordinary positive
resistance a current flows in at the terminal connected to the
positive pole of the source and out at the other terminal. This direc-
tion of current flow is considered positive and the value of the resistance
R, in ohms is given by Ohm's law as R = E/I where E is the applied
voltage and / is the current in amperes. Similarly a definite current /
may be passed through the resistance and a potential difference or drop
E = RI will appear across its terminals. With positive resistances it
makes no difference whether we "apply an e.m.f." or "pass a current".
The resistance may be a very simple device such as a coil of wire which
absorbs energy from the circuit at a rate W = EI — PR watts.
It is possible, however, to construct assemblages of apparatus which
have the property of keeping the ratio of the voltage across a pair of
terminals to the current at the terminals constant, but with the relative
direction of the voltage and current opposite to that which a positive
resistance would give. In such devices the resistance is negative and
the apparatus contributes power to the circuit with which it is con-
nected. Each such device necessarily includes a source of energy such
as a battery and some means such as a vacuum tube for controlling the
delivery of this energy to the circuit. There are two varieties of such
devices. In one case, the internal arrangement of the mechanism is
such that, if a definite voltage is applied to the terminals, a current
flows in a direction opposite to the applied e.m.f. In the other, if a
definite current is passed through the system, the drop across the
terminals will be opposite in direction to that caused by a positive
resistance. These two arrangements are essentially different and
cannot be used interchangeably in a given circuit, though either one can
give any desired value of negative resistance. If the wrong arrange-
ment is used instability or singing will occur. To know whether a
given negative resistance will work satisfactorily in a given circuit it is
not sufficient to know its value in ohms. Something must be known
485
486
BELL SYSTEM TECHNICAL JOURNAL
about its Internal arrangement and about the impedance of the circuit
in which it is to work.
Regenerative Negative Resistances
One of the simplest ways to produce a negative resistance is to inter-
connect the input and output terminals of a one-way amplifier. This
gives a regenerative arrangement because part of the output energy
of the amplifier is fed back into the input circuit. The type of negative
resistance obtained depends upon the way in which the interconnection
is made.
Fig. 1 — Ideal one-way amplifier.
Fig. 1 shows schematically an ideal one-way amplifier for this pur-
pose. It has a pair of input terminals 1, 2, and a pair of output ter-
minals 3, 4. The impedances between the input and output terminals
are pure resistances Ri and R^, respectively. Some mechanism, indi-
cated symbolically by the arrow, is provided, which produces an e.m.f.
in the output circuit which is proportional to the input current. The
nature of this mechanism is not of importance to this discussion except
that it is a one-way device. The mutual impedance M is the ratio of
the e.m.f. generated in the output circuit to the current in the input
circuit. This ratio may be adjusted by suitable means such as a
potentiometer but is otherwise constant and includes no phase shift.
The internal connections are assumed to be such that when the input
terminal 1 is positive to 2 the e.m.f. in the output circuit tends to make
terminal 3 positive with respect to 4.
Series Negative Resistance
In Fig. 2 the input and output circuits of the ideal amplifier are
connected in series with each other to a source of e.m.f. E and a re-
Ro
^8
Fig. 2 — -One-way amplifier connected as a series negative resistance.
NEGATIVE IMPEDANCES AND THE TWIN 21-TYPE REPEATER 487
sistance Rq in such fashion that the e.m.f. in the output circuit of the
ampHfier tends to increase the current. Assume now that the e.m.f.
E is applied and a current /o flows in the series circuit.
E + {M - Ro- Ri - Ro)Io = 0. (1)
The drop across the ampUfier is:
e= {R, + R,- M)h, (2)
and the net resistance of the whole amplifier is:
r = y=R, + R,- M. (3)
â– i-o
It may aid in understanding the behavior of this system to assume,
first, that M is zero so that the circuit consists simply of the three
positive resistances Ro, Ri and Rz in series and then consider what
happens as M is gradually increased. The e.m.f. appearing in the
output circuit of the amplifier acts to reduce the drop e across the
terminals 1, 3 and to increase the current Iq. The e.m.f. E must be
reduced if the current is to be kept constant. The curves of Fig. 3
show how the resistances and current vary as ^1/ changes, E being
constant.
When M = Ri -{- R2 the drop e and the resistance r become zero.
The amplifier then ceases to take power from the circuit and supplies
its own losses. If this condition could be exactly obtained the ter-
minals 1, 3 might be short-circuited and the e.m.f. E removed, without
changing the current which would continue to flow in the amplifier.
If, however, the e.m.f. were removed or the circuit opened without
short-circuiting the terminals of the amplifier the current in the input
circuit, and, consequently, the e.m.f. in the output circuit of the
amplifier would disappear and the system would become inactive.
If, now, M is further increased so that it approaches i<!o + -Ki + R-i
the current increases indefinitely, or the e.m.f. E required to sustain
the current at a given value approaches zero. Under these conditions
the drop e and the resistance r become negative and the amplifier
supplies not only its own losses but also part of the energy dissipated
by the resistance Ro- It does so under the control of the e.m.f. E,
however, and if this e.m.f. is removed the system becomes inactive as
before. At the limit when M = Ro -\- Ri -{- R^, the amplifier supplies
all the losses in the system and any current /o, once started, continues
indefinitely.
This ideal condition is not realized in practice. Either M is slightly
too small, in which case the current decreases when E is removed, or it
488 BELL SYSTEM TECHNICAL JOURNAL
is too large so that any value of E however small starts a current which
thereafter increases because the amplifier supplies more than enough
energy to sustain the current. This increase continues until checked
by the inability of the amplifier to deal with larger currents. In effect
M is reduced to the point where r is again equal to — i?o, after which the
current continues at a constant value.
The arrangement shown in Fig. 2 can therefore be made to provide
any negative resistance between r — and r = — Rq without causing
instability or a tendency to sing. Such a system is stable when the
algebraic sum of all the resistances in series in the circuit is positive.
This behavior is typical of a large number of arrangements that are
able to furnish negative resistances. All such arrangements will be
referred to as series negative resistances to distinguish them from another
type which will be described below.
It should be noted that if the sign of M is reversed, for example, by
interchanging the two wires connected with the output terminals 3, 4,
no negative resistance results. As M increases, the current /o de-
creases, or the e.m.f. E must be increased to maintain the current,
but no matter how large AI is made, the direction of the drop e and sign
of the resistance r do not change though the latter approaches <».
The Unstable Condition
So far nothing has been said as to the nature of the e.m.f. E. In the
ideal case, when the system is stable, the current wave is a copy of the
voltage wave as in any circuit having a pure resistance. What hap-
pens when the circuit is unstable depends upon the nature of the ampli-
fier or other device used to produce the negative resistance and not
upon the e.m.f. E. This may be of any kind and of minute size, such
as that resulting from thermal agitation in the resistances forming
part of the apparatus. If the amplifier is able to amplify direct cur-
rents, the resulting disturbance may be a direct current limited only
by the ability of the apparatus to supply energy to the circuit. Where
transformers, condensers, etc., are involved the disturbance settles
down to an alternating current which may contain many harmonics
or may be almost a pure sine wave. These effects are called "sing-
ing." The final frecjuency, amplitude and wave shape depend upon the
makeup of the apparatus in a way which is beyond the scope of this
paper.
Shunt Negative Resistance
By connecting the terminals of the ideal one-way amplifier in parallel
as shown in Fig. 4, a negative resistance will be obtained which is
typical of the second type or shunt negative resistance.
NEGATIVE IMPEDANCES AND THE TWIN 21-TYPE REPEATER 489
Referring to Fig. 4, the current in the input circuit of the ampHfier is:
/i
Ri
(4)
^
f
^
P
SERIES NEGATIVE
RESISTANCE
(4.
i
f
\
1
V//
^
//y
>
^
^
i
^
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r
i
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\
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\
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^
^
^
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v^
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y//.
1
"^
c
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\
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^
/
^
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\
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^
^
^
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^
^
^
i
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of
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y
>is
y/A
iMSTA
'///
/^
lo
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^
^
^
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f^z
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\
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gN
P AB.
ORBS
ENE
J.
■« EMf
rs
v//
^
^
ENE
^GY
^
i^/
^
W/
^
^
Fig. 3 — Curves illustrating properties of series negative resistances.
The current in the output circuit is:
e - Mh Ri - M
h = B = n D g.
R2
R1R2
(5)
490
BELL SYSTEM TECHNICAL JOURNAL
and the current in the main circuit is:
T P- - g T _L T ^1 + ^2 - M
h = S = /l + /2 = ^5-n «.
i?o
from which
RiRi
e
r = -v
R1R2
h R1 + R2-M'
and the appHed voltage E is:
E ^ Io{Rq + r) =
(6)
(7)
(8)
Fig. 4 — ^One-way amplifier connected as a shunt negative resistance.
With this arrangement, the e.m.f. generated in the output circuit of
the ampUfier opposes the current I2 due to the e.m.f. E, and as M
increases, the current /o in the main circuit decreases and the resistance
of the ampUfier increases. The curves of Fig. 5 show how the resist-
ances and current vary as M changes, E being constant. To keep /o
constant, it would now be necessary to increase E.
When M = Ri the current I2 becomes zero.
When M = i?i + i?2 the current /o falls to zero, the potential
e = E, the current h has reversed in direction, the resistance r = <»
and the amplifier just supplies its own losses. If the circuit outside
the amplifier is now opened, the condition of the amplifier is the same
as when the short circuit was applied to Fig. 2 and the current circulat-
ing in the amplifier will continue. If E is removed without opening
the circuit, Ro will draw energy from the amplifier, thus reducing /i and
causing all currents and voltages to disappear. The amplifier is still
under the control of the e.m.f. E.
For the arrangement of Fig. 4 to become unstable it is necessary for
the amplifier to maintain or increase the voltage e after the controlling
e.m.f. E is removed. For the amplifier to maintain the voltage e it is
necessary that:
RoRi
^-r/'
Ro + R,
RoR\
Ro + R^
(9)
+ i?2
NEGATIVE IMPEDANCES AND THE TWIN 21-TYFE REPEATER 491
from which
and from (7),
Hence if
M = R, + R,+
r = -Rn
R,Ro
R, + Ro < M <Ri+ R2 +
R,Ro
R,
(10)
(11)
(12)
the impedance r is a negative resistance greater in magnitude than Ro but
the system cannot sing because the ampUfier cannot maintain or increase
Fig. 5 — Curves illustrating properties of shunt negative resistances.
492 BELL SYSTEM TECHNICAL JOURNAL
the voltage e after E is removed, even though the current /o flows
against E and the source is receiving energy from the amplifier.
If M becomes greater than the upper limit given by equation (10)
the system passes out of control by the e.m.f. E and becomes unstable
or sings. By short-circuiting the terminals 1, 2, it would be possible to
increase M until it is greater than the value given by equation (10)
which would make r numerically smaller than Rq. On removing the
short circuit, however, a disturbance would begin and grow until
checked by the limitations of the amplifier so that, in effect, M would
be reduced and r again made equal to —.Ro-
lf M is reversed in sign, for example, by interchanging the two wires
connected to the output terminals 3, 4, no negative resistance results.
As M increases, the current In increases. The resistance r decreases,
approaching zero as M becomes indefinitely great.
From these facts it is seen that a negative resistance of any desired
value may be inserted in a circuit having any positive resistance Ra
provided that the inserted resistance has the characteristics of the
series type when the inserted negative resistance is numerically smaller
than the positive resistance or the characteristics of the shunt type
when the negative resistance is numerically larger than the positive
resistance.
Other Forms of Negative Resistance
All known devices for producing negative resistance fall into one
or the other of the two classes described above.
Arrangements are known which exhibit one type of negative re-
sistance at one pair of terminals and the other type at a different pair
but not both types at the same pair of terminals at the same time.
Certain apparatus involving gaseous conduction or electronic dis-
charge exhibit negative resistance effects. Fig. 6, for example, shows
an arc burning between two electrodes which are connected in series
with a resistance and inductance serving as ballast to a source of d-c.
power. The ballast serves to stabilize the arc and hold the current
drawn from the source constant and also to prevent the passage of
alternating current through the source from the arc. The arc has a
positive resistance with respect to the d-c. circuit, since it consumes
d-c. power, but this resistance varies with the current in such a way
that an increase of current is accompanied by a reduction of the po-
tential drop across the arc.
If an alternating current is superimposed upon the direct current
through the arc by means of the taps a and h it encounters a negative
resistance. If a circuit consisting of a resistance R, inductance L and
NEGATIVE IMPEDANCES AND THE TWIN 21 -TYPE REPEATER 493
condenser C is l)ridged across the arc as shown and the resistance is
made large, nothing occurs, but if the resistance is reduced to a certain
critical value a state of oscillation is established. This oscillation
causes an alternating current to flow through the resonant circuit and
the arc. If the oscillation is of audible frequency the arc will emit a
BALLAST
<nnnrinrb — °^wamv°
ARC
SOURCE OF
POWER
Fig. 6 — -Arc as a negative resistance.
singing or whistling sound. This property of the arc has found useful
application as a generator of high-frequency oscillations in the Poulsen
arc used in radiotelegraphy. The negative resistance of the arc has
series characteristics as oscillations will not occur if there is an excess
of positive resistance in the oscillating circuit.
The dynatron,^ on the other hand, has shunt characteristics as it is
unstable when the external resistance is made large.
Negative Resistances of the Ideal 21-Type Circuit
Fig. 7 shows the ideal one-way amplifier of Fig. 1 connected with an
ideal hybrid coil to form a 21-type repeater circuit. The ideal hybrid
reversing
switch
jisumsm.
Ifty*
o-^7Rnnr-
-nr^wp-
o— nnnsT' — l — i^tpsw^
li —
Fig. 7 — Ideal 21-type circuit.
coil is assumed to have windings of zero resistance, no leakage react-
ance, no capacitance in or between the windings, no core loss and neg-
ligible exciting current.
^See "The Dynatron," by A. \V. Hull, Proc. I. R. E., February, 1918.
494 BELL SYSTEM TECHNICAL JOURNAL
This 21 -type circuit is connected between two equal resistances, Rq.
Ri = 3^i?o and Ri = Ro, assuming that the hybrid coil is designed for
equal impedances at the two pairs of line terminals and the drop
terminals. If an e.m.f. E acts in series with the resistance Ro at the
left side of the repeater and the mutual impedance AI of the amplifier
is zero, a current,
/. = 2i. (13)
flows at the left hand terminals 5, 8 of the hybrid coil and in the input
circuit of the amplifier. One-half of the power entering the repeater
is absorbed in the input resistance Ri of the amplifier and the other
half is absorbed in the output impedance i?2- In accordance with a
well known property of the hybrid coil, no current will flow in the
right-hand resistance Ro. At a given instant this input current may
be assumed to have the direction indicated by the short arrows /i.
By increasing M the amplifier can be made active, causing an amplified
current /o to flow in series through the line windings of the hybrid coil
and the connected resistances Ro. By throwing the reversing switch to
one side I^ may be made to flow in the same direction as /i at the ter-
minals 5, 8 as indicated by the long arrows marked I2. For con-
venience, this will be referred to as the "direct connection." Changing
the reversing switch changes the direction of I2 with respect to /i,
giving the "reverse connection." As the hybrid coil is balanced, the
output power of the amplifier does not react upon the input circuit.
Putting A for the amplifying ratio of the 21 -type circuit,
^ = ^; • (14)
The total current flowing at the terminals 5, 8 is:
/o= /l+/2 = ^(l+^), (15)
ZKo
and the active resistance of the 21 -type circuit is:
As the amplification is increased, the current Iq increases while r falls
to zero and becomes negative, thus exhibiting series characteristics.
If A is increased without limit, r approaches —Rq in magnitude but
NEGATIVE IMPEDANCES AND THE TWIN 21-TYPE REPEATER 495
cannot reach it, while A remains finite. That is, the system shown in
Fig. 7 cannot sing. This is also obvious from the fact that the hybrid
coil is balanced. However, the resistance r does not depend upon
holding Rq at the terminals 5, 8 constant. If the resistance at the
terminals 5, 8 is reduced to a lower value Rq , while that at the terminals
6, 7 is held constant at i?o, the output energy of the amplifier is per-
mitted to reach the input terminals 1, 2 and when —Rq = r instability
or singing can occur.
Throwing the reversing switch to give the reversed connection has
the effect of reversing the sign of the amplification A. The total
current Jo at the terminals 5, 8 decreases to zero, reverses and increases
as A increases, while r increases, passes discontinuously from + oo to
— CO and decreases in magnitude. Again r approaches — i?o as A
increases indefinitely, but cannot reach it. However, by increasing
the resistance connected to the terminals 5, 8 to a higher value R^'
such that —Rq — r, instability will occur. The reversed connection
thus gives a negative resistance of shunt characteristics.
Referring to Fig. 7 and assuming that the switch is thrown to give
the directions of current flow indicated by the arrows, transfer the
e.m.f. E to the right-hand end of the diagram. This change will not
change the direction of /i in the input circuit of the amplifier or the
direction h at any point. The current /i will now be found at ter-
minals 6, 7 instead of 5, 8 and will be flowing in the direction opposite
to h. From this it will be seen that a 21-type circuit which is direct-
connected with reference to terminals 5, 8, giving a series type negative
resistance, will be reverse-connected, and give a shunt type negative
resistance at the opposite terminals 6, 7. Changing the reversing
switch reverses the conditions at both pairs of terminals.
Non-Ideal Devices
The discussion has so far been confined principally to certain ideal
conditions which can only be approximated in practice, but considera-
tion of these simple cases will serve to illustrate the important funda-
mental properties of negative resistances and the requirements that
must be met to insure stable operation.
To obtain a pure negative resistance from a one-way amplifier or
from a 21-type repeater circuit requires that there shall be no phase
shift in the process of amplification. This can only be approximated in
practice because even a resistance coupled amplifier system involves
small inductances and capacitances in the tubes and wiring which pro-
duce phase shifts at high frequencies. Commercially practicable trans-
496
BELL SYSTEM TECHNICAL JOURNAL
formers, choke coils, and condensers whicli are so useful in assemblages
of apparatus which include vacuum tubes further limit the range of
frequency over which an approximately pure negative resistance may
be obtained. In some cases, this may not be a serious disadvantage.
Suppose, for example, it is desired to reduce the effective resistance of
a series resonant circuit in order to obtain more nearly ideal perform-
ance at the resonant frequency. It would be sufficient to arrange a
negative resistance in series with the resonant circuit which would
produce the desired result at and near the resonant frequency and
which would produce no harmful effect at other frequencies even though
it departed widely from the value at the resonant frequency. In
other cases the variation of the negative resistance with frequency and
the introduction of reactive components do no serious harm and may
even be quite useful as in the case of the twin 21 -type repeater to be
described below. In still other cases the difficulties of producing a
negative resistance of satisfactory characteristics may be very great.
General Negative Impedance
The arrangements described above produce under ideal conditions
pure negative resistances.
It has been shown by R. C. Mathes and H. W. Dudley that it is
possible to produce any desired negative impedance provided that the
positive of this impedance can be constructed in the form of a network.
e '^z-
Fig. 8 — Series type negative impedance.
Fig. 8 shows in simplified form the arrangement invented by Mathes,
and Fig. 9 shows the arrangement due to Dudley. Each of these
arrangements requires a distortionless one-way amplifier whose input
impedance (terminals 1, 2) is substantially infinite. This condition is
easily approximated by using vacuum tubes. In discussing the
behavior of such arrangements, it is necessary to use the ratio, AIv, of
NEGATIVE IMPEDANCES AND THE TWIN 21-TYPE REPEATER 497
the e.m.f. generated in the output circuit of the ampHfier to the voltage
impressed on its input terminals, instead of the mutual impedance of
the amplifier, because the input current is negligibly small. This
ratio may be adjusted by some suitable means such as a potentiometer.
Referring to Fig. 8, let Z be the positive of any desired negative
impedance such that a network having the impedance, Z.v = Z/M^ — 1,
may be constructed of physically available parts, M^ being a real
Fig. 9 — Shunt type negative impedance.
number greater than 1. Rk = Ro/My — 1 is a pure positive resistance.
Next assume that a current, /, is flowing through the circuit between
terminals 5 and 6. The e.m.f. generated in the output circuit of the
amplifier is {Rs + Z\)IMy. It acts in the direction which tends to
increase the current. The voltage e required at the terminals 5, 6 to
produce this current is, then.
e = {Ry + Zv + R2)I - (Rx + Zx)IM,,
from which the impedance Zz is:
-z,
(17)
(18)
which is the desired negative impedance. Due to the arrangement of
the circuit this impedance has series characteristics.
Referring to Fig. 9, Z,v is a positive network. Assuming that an
e.m.f. e is applied to the terminals 7, 8, the e.m.f. generated in the
output circuit of the amplifier is eM,, which acts in opposition to e to
reduce or reverse the current. The current at the terminals 7, 8 is,
then,
_ e — eM„
i?2 + Zat '
(19)
498 BELL SYSTEM TECHNICAL JOURNAL
and the impedance Zi at the terminals 7, 8 is:
^â– ' = 7 = A - ^' (20)
which consists of the desired negative impedance —Z and a negative
resistance if Mv > 1- By connecting the positive resistance,
7?3 = R^/Mv — 1, in series with Z/ this negative resistance is neu-
traUzed and the desired negative impedance is found between the
terminals 8 and 9. This impedance has shunt characteristics.
In both of these arrangements it is possible, without changing the
constants of the network Z.v, to give the negative impedance any
desired magnitude by adjusting the value of AIv and making the cor-
responding change in the resistance Rn or R3.
Boosters
The name "booster" has been applied to a negative impedance of
suitable characteristics connected in series with or bridged across a
telephone circuit in order to introduce energy when a wave passes and
so produce a transmission gain. Such devices have certain interesting
theoretical properties.
Series Booster
Fig. 10 shows an impedance Zs connected in series between the two
parts of a telephone line having the characteristic impedance Zq.
Zs
^/MAAAMp
Fig. 10 — The series booster.
Assume first that Zs is a positive impedance having the same angle
as Zo and that a wave is traveling over the line, for example, from left
to right. The effect of the inserted impedance is to reduce the current
in the line wires at the point of insertion, weakening the wave that
passes on to the receiver and causing a reflected wave to return to the
source. The transmission loss^ caused by the inserted impedance is:
L= 20 1og:o(l+;^J, (21)
"The values of losses, return losses and gains will be e.\pressed in decibels (db)
throughout this paper.
NEGATIVE IMPEDANCES AND THE TWIN 21-TYPE REPEATER 499
and the return loss ^ due to the irregularity is:
5= 20 1ogio(l + ~) • (22)
If, now, Zs is made a negative impedance of the series type smaller
in magnitude than 2Zo, the potential difference between its terminals
reverses in sign, the current at the point of insertion increases, the loss
becomes a gain and the reflected wave reverses in sign. As Zg ap-
proaches — 2Zo, the transmitted and reflected waves increase until
singing occurs; but the reflected wave is always smaller than the
transmitted though they approach each other as the gain increases.
Such a booster, therefore, causes a smaller returned wave or echo than
an ideal 21 -type repeater circuit working between ideal line impedances
which always returns a wave toward the source which is equal to that
transmitted toward the receiver.
The series booster would also operate if Z^ were made a shunt type
negative impedance greater in magnitude than 2Zo, but in this case the
current at the booster and the wa\-e traveling toward the receiver
would be re\ersed in phase and the reflected wave or echo would be
greater than the wave traveling toward the receiver. This arrange-
ment would, therefore, give greater echoes for a given gain than a 21-
type repeater. The curves of Fig. 12 show the relation between the
return loss and transmission gain for these boosters in comparison with
a 21 -type repeater.
The echoes referred to above are, of course, those inherent in the
operation of the devices described and would not occur if a 22-type
repeater were used with perfect lines. Echoes due to line irregularities
would be amplified to the same extent by boosters as by any other
type of two-way repeater giving the same gain.
Shunt Booster
Fig. 11 show^s an impedance Zb bridged across the line. The effect
of this impedance is to reduce the wave tra\eling toward the receiver,
causing a transmission loss,
L= 20 1og:o(l+^J, (23)
and causing a reflected wave to return to the source with a return loss,
5= 20 1ogio(l+^) • (24)
^ When a wave is partially reflected at an irregularity the relation between the
reflected part and the original wave, expressed in decibels, is called the return loss.
500
BELL SYSTEM TECHNICAL JOURNAL
In this case the current in the line leading toward the source is in-
creased; that is, the reflected wave is of opposite phase to that reflected
by an impedance in series with the line.
Fig. 11 — ^The shunt booster.
If Zb is made a negative impedance with shunt characteristics and
greater in magnitude than Zo/2, the current through Zh reverses in
sign, the wave transmitted toward the receiver increases, the transmis-
20
'o
I
o
/
^
/
//
/
z
/
/
/
<
o
y
/
/
/
z
a.
^ 10
u
a.
y
/
/
Y
/
A/
-^
/
/
/
J
^
â– ^
b/
/
/
i
oj
/
/
A
)
G
mn/
db
10
1
2
/
/
CURVE A — SHUNT TYPE
/
/
NEGATIVE IMPEDANCE
IN SERIES WITH THE
/
LINE OR SERIES TYPE
NEGATIVE IMPEDANCE
10
if)
/
BRIDGEDON THE LINE.
/
CURVE B— 21 TYPE RE-
PEATER.
3
/
CURVE C— SERIES TYPE
z
u
/
IMPEDANCE IN SERIES
WITH THE LINE OR
SHUNT TYPE IMPED-
ANCE BRIDGED ON THE
20
LINE.
Fig. 12 — Echoes caused by boosters and 21-type repeaters
sion loss becoming a gain and the reflected wave reverses in sign, thus
reducing or reversing the current in the line leading toward the source.
The relation between the magnitude of the echo and the gain is the
NEGATIVE IMPEDANCES AND THE TWIN 21-TYPE REPEATER 501
same as for the series type booster described above except that the
reflected waves are opposite in phase. This makes it possible to elimi-
nate the echo by combining two boosters in one repeating device as
described below.
The shunt booster would also operate if Zj, were made a series type
negative impedance smaller in magnitude than Zo/2, but in this case
the wave traveling toward the receiver would be reversed in phase and
the echo wave would be greater than the transmitted wave.
Singing Points of Various Forms of Repeaters
When a line of characteristic impedance Zo has a certain return loss
Sh its impedance will lie between a maximum value of mZo and a
minimum of Z^/m where
Si = 20 logio "^^ • (25)
m — \
If two pieces of such a line are joined through a repeating device the
high and low impedances may combine in three different ways which
give the greatest tendency to sing with different types of apparatus.
The series type negative impedance, whether connected in series
with or across the line, has the greatest tendency to sing when the
minimum impedances of both lines occur at the same frequency and the
shunt type negative impedance has the greatest tendency to sing when
the maximum impedances occur at the same frequency. The 21 -type
repeater has the greatest tendency to sing when the maximum im-
pedance of one line and the minimum of the other occur at the same
frequency, the internal connections of the repeater determining which
impedance must be high. In the 22-type repeater any of these com-
binations may be the worst, depending upon the internal arrangement
of the repeater circuit.
The series booster (with series type negative impedance) will sing
when
Z. + ^ = 0. (26)
m
Substituting Zs obtained from this relation in equation (21) and
remembering that the loss L becomes a gain G\ when Zs is negative,
the gain which will produce singing is:
Gs= 20 logic (l -I)' (27)
502 BELL SYSTEM TECHNICAL JOURNAL
This gain is. of course, the gain which a booster having the impedance
Z, obtained from equation (26) would produce when connected be-
tween two impedances Zq. The actual gain of the booster, like that
of any other t>'pe of repeater approaches infinity as the singing condi-
tion is approached.
The shunt booster (with shunt type negative impedance) will sing
when
Zft + ^ = 0. (28)
Substituting the value of Zb from this equation in equation (23)
shows that the relation given in equation (27) also holds for the shunt
type booster.
It is well known that when a 22-type repeater giving the gain 6*22 in
each direction is connected between two lines having the return loss Si
singing will occur when
G21 = Si, (29)
if the worst combination of unbalances occurs.
It is also well known that under similar conditions the gain of a
21 -type repeater is:
Go, ^ Si- 6db, (30)
because of the fact that waves reflected from the irregularities in both
lines combine in the input circuit of the amplifier.
The curves of Fig. 13 show the singing gain as a function of line
return loss for boosters, 21-type and 22-type repeaters. These curves
together with the curves of Fig. 10 indicate that ideal boosters con-
sisting of series type negative impedances in series with the line or
shunt type negative impedances bridged across the line have properties
intermediate between those of 21 and 22-type repeaters with respect to
the amount of echo and margin against singing for a given transmission
gain. These properties are particularly favorable at low gains.
In practice, however, it is usually necessary to limit the amplification
to a definite band of frequencies in order to avoid the effect of imped-
ance unbalances and interfering disturbances at frequencies outside
these limits. This must be accomplished by the use of inductance
and capacitance in the form of filters, transformers, choke coils or
condensers. It is also desirable to couple the series booster to the line
by means of a transformer having two equal windings, one in each line
conductor, to enable one booster mechanism to operate without
unbalancing the line and to permit the passage of low frequency signal-
ing waves from one part of the line to the other without interference
NEGATIVE IMPEDANCES AND THE TWIN 21-TYPE REPEATER 503
from the booster. For similar reasons, condensers must be connected
in series with the shunt booster when it is bridged on the line. These
devices, particularly the filters, shift the phase of the amplified waves,
and modify the negative impedances so that the gain varies with
frequency in the useful range to a greater extent than is the case with
the 21 and 22-type repeaters and the echoes are increased. This
variation of gain is due to the fact that the booster, in effect, super-
imposes an amplified wave upon the wave that would exist if the
30
20
<
o
o
z
o
S 10
in
/
/
-^
/
A
#
/
«<t
r-
/
/^
^-^/^
^
%<<^
^
S
/
4
^'
/
}
/
/
'/
/
/
/
/
y
y
/
/
r
/
/
^
/
/
1
TURf
NJ LC
)SS
db
2
/
/
Fig. 13 — Singing gains of boosters and repeaters.
booster were removed. The received wave, being the resultant of
these two waves, varies with the phase angle between them.
It should also be noted that boosters do not avoid the problem of
matching line impedances or the difficulties due to impedance irregu-
larities in the line. To obtain a gain that is constant over a wide
range of frequencies, the negative impedance must be fitted to the
line impedance over this range and there must be no large irregularities.
It will be shown below that most of the difficulties described above may
be avoided by using a series and a shunt booster in combination.
504
BELL SYSTEM TECHNICAL JOURNAL
Negative Impedances Arranged in T or IT Networks
It has been pointed out by G. A. Campbell, H. Mouradian/ and
possibly by others, that three negative impedances can be grouped into
a r or a TT network which may be inserted in a telephone line. Such
a network is able to amplify waves traversing the line without causing
echoes if the values of the impedances are suitably chosen. In order
to avoid singing, the impedances in series with the line must be of the
series type, and those bridged across the line, of the shunt type.
A Double Booster
Fig. 14 shows a network of impedances connected between two pieces
of telephone line having the characteristic impedance Zq. These lines
are assumed at first to be free from irregularities. The branches
ac and he are fixed networks, each having the impedance Zq. Branches
ah and cd are networks whose impedances can be varied reciprocally
from the value Zo, that is, if one impedance is multiplied by a factor
pZo
LINEW
LINEE
Fig. 14 — Double booster.
p, the other is divided by the same factor. The factor p may be
positive or negative, and may be complex. Branches ah, ac, cd and the
line E may be considered as forming the arms of a Wheatstone bridge,
of which the branch he is one diagonal and the line W is the other.
This bridge is balanced; consequently, the impedance connected to the
line W consists of two parallel circuits, one comprising the branch ah
in series with the line E and the other comprising the branches ac
and cd in series. This impedance is independent of p, being equal to
Zo- By symmetry, the impedance connected to the line E is also equal
to Zo, so no reflection occurs at the terminals of the network.
Assuming that a wave arrives, for example, over the line W and is
*"Long Distance Transmission Problems," by 11. Mouradian, Journal of the
Franklin Instilute, \'o\. 207, No. 2, February, 1929.
NEGATIVE IMPEDANCES AND THE TWIN 2 1 -TYPE REPEATER 505
transmitted to the line E, the ratio of the voltage across the terminals
a, d to that impressed on the hne £ is (1 + p)/l and the transmission
loss through the network is:
L = 20 login (1 + p). (31)
This loss becomes a gain when p becomes negative and the network
acts as an amplifier.
Examination of Fig. 14 shows that the branches ab and cd are each
connected to a constant impedance Zo, hence, if p lies between and — 1 ,
the branch ab must be a series type negative impedance and cd of the
shunt type. If p lies between —1 and 
