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ELECTRICAL    ENGINEERS'    HANDBOOK 

Electric    Communication 
and    Electronics 


WILEY    ENGINEERINGS   HANDBOOK   SERIES 


OF   ENGINEERING    FUNDA- 

MENTALS.     Edited  by  OVID  W.  ESHBACBE. 

KENT'S   MECHANICAL  ENGINEERS*   HAND- 
BOOK.     Twelfth  Edition. 

DESIGN    AND    PRODUCTION.      Edited   by 
COLIJN" 


PO^VER.       Edited  by  J.   K. 

ELECTRICAL    ENOINEERS* 

Fourth.  Edition. 

ELECTRIC     POWER.       Edited     by 
and  WILLIAM:  A.  DEL 


ELECTRIC     COMMUNICATION    AND 
ELECTRONICS.     Edited  by  HAJ^OLT* 
and   KLrsrox 


ENGINEERS^    HANDBOOJEC. 

Edition.      Edited  by  tne  late  ROBERT  PEELE. 
HANI>KOOiC    OF    MIJNERAL    ORESSKNTG. 

ORES    AND    INDUSTRIAL    MINERALS. 
Edited  by  AetTHxna  F. 


ELECTRICAL 

HANDBOOK 

Electric  Communication 
and  Electronics 


Prepared  by  a  Staff  of  Specialists 

HAROLD  FENDER,  PH.D.,  Sc.D. 

and 

KNOX  McILWAIN,  B.S.,  E.E. 

Editors 


FOURTH  EDITION 


WILEY  ENGINEERING 
HANDBOOK  SERIES 

NEW   YORK    :    JOHN    WILEY    &    SONS,    ING. 
LONDON     :    CHAPMAN    &    HALL,    LIMITED 


COPYRIGHT  1914,   1922,   1936,    195O 

BY 
JOHGNT     "WlLEY     &    SoisTS,     IlSTC- 


Copyright  1914  renewed,   1941 

Copyright  1922  renewed,   1949 

3By  John.  Wiley  &  Sons,  Inc. 


Copyright,  Canada,    1936,   195O;  International  Copyright,   1936,  195O 
John  Wiley  &  Sons,  Inc.,  Proprietors 


Rights  Reserved 

fta  l>ook  or  curby  yya,rt  thereof  must  not 
be     reproduced,    'in     any    form, 
the  written  %>ermi>&afion  of  the 


PRI3STTED     I3ST     THE    UNITED     STATES     OF    AI^ERICA 


PREFACE 

The  first  edition  of  Fender's  Handbook  for  Electrical  Engineers,  compiled  by  a  staff  of 
specialists  under  the  editorship  of  Harold  Fender,  appeared  in  1914.  The  second  edition, 
under  the  joint  editorship  of  Fender  and  William  A.  Del  Mar,  was  published  in  1922- 
Both  these  editions  covered  all  branches  of  electrical  engineering  as  well  as  a  large  amount 
of  material  dealing  with  allied  fields  of  interest  to  electrical  engineers. 

The  third  edition,  published  in  1936,  was  divided  into  two  volumes:  one  on  electric 
power  under  the  editorship  of  Fender,  Del  Mar,  and  Knox  Mcllwain;  the  other  on  elec- 
trical communication  and  electronics  under  the  editorship  of  Fender  and  Mcllwain. 
Certain  tables  and  fundamental  theory  were  duplicated  in  the  two  volumes  in  order  that 
each  might  be  complete  and  independent  of  the  other. 

This  plan  met  with  such  enthusiastic  response  that  it  has  been  continued  in  the  fourth 
edition.  The  growth  of  knowledge  and  the  greater  degree  of  specialization  in  the  various 
phases  of  electrical  engineering  have  necessitated  a  considerable  enlargement  of  both 
volumes.  Careful  selection  and  Compression  have  been  exercised  in  an  effort  to  keep 
the  books  compact  and  readable.  The  treatment  of  subjects  of  decreased  importance 
and  those  which  are  adequately  treated  by  other  handbooks  of  the  Wiley  Handbook 
Series  has  been  either  curtailed  or  left  unchanged  in  length. 

The  bibliographies  have  been  prepared  with  the  idea  of  assisting  the  reader  to  further 
study  of  each  subject-,  and  they  reflect  each  author's  idea  of  this  plan.  The  publications 
referred  to  are,  in  general,  in  the  Engineering  Societies  Library,  29  West  39th  Street, 
New  York,  N.  Y.  Most  of  them  may  be  borrowed  from  the  Library  by  members  of  its 
Founder  Societies,  the  American  Society  of  Civil  Engineers,  American  Institute  of  Mining 
and  Metallurgical  Engineers,  American  Society  of  Mechanical  Engineers,  and  American 
Institute  of  Electrical  Engineers. 

Seventy-eight  specialists  in  their  respective  fields  have  contributed  to  this  fourth  and 
entirely  rewritten  edition  of  the, Electronics  and  Communication  portion  of  the  Electrical 
Engineers1  Handbook,  as  compared  with  twenty-seven,  forty-five,  and  fifty-seven  in  pre- 
vious editions.  This  reflects  the  rapid  widening  in  the  electronics  field.  In  particular, 
frequency  modulation  and  all  the  pulse  techniques  in  both  the  communication  and  radar 
fields  appear  in  the  volume  for  the  first  time.  The  increased  complexity  and  importance 
of  radio  aids  to  navigation  are  also  of  interest. 

The  editors'  thanks  are  due  to  the  many  well-known  and  busy  men  who  have  con- 
tributed textual  material,  both  for  their  unselfish  efforts  to  make  this  a  reliable  reference 
work  and  for  their  continued  patience  with  editorial  vagaries.  They  are  also  due  to 
Messrs.  R.  L.  Jones,  R.  K.  Honaman,  and  A.  R,  Thompson  of  the  Bell  Telephone  Labora- 
tories, Mr.  Frank  A.  Cowan  of  the  American  Telephone  and  Telegraph  Company,  and 
Mr.  E.  W.  Engstrom  of  RCA  Laboratories  for  aid  in  the  organization  of  the  book. 

HAEOLD  FENDER 

KNOX   MclLWAIN 


LIST  OF  CONTRIBUTORS 

Maurice  Artzt,  Research  Engineer,  Radio  Corp.  of  America,  RCA  Laboratories  Divi- 
sion. Facsimile. 

W.  F.  Bailey,  Sr.  Television  Engineer,  Hazeltine  Electronics  Corp.  Television  Receivers. 
^  Loy  E.  Barton,  Research  Engineer,  Radio  Corp.  of  America,  RCA  Laboratories  Divi- 
sion. Amplifiers. 

Dr.  R.  M.  Bozorth,  Research  Physicist,  Bell  Telephone  Laboratories.  Magnetic  Mate- 
rials. 

Dr.  J.  G.  Brainerd,  Professor,  Moore  School  of  Electrical  Engineering,  University  of 
Pennsylvania.  Units  and  Conversion  Factors,  Symbols  and  Abbreviations. 

R.  B.  J.  Bnmn,  Sr.  Engineer,  Hazeltine  Electronics  Corp.     Television  Receivers. 

T.  J.  Buzalski,  Television  Station  Engineer,  National  Broadcasting  Company.  Tele- 
vision Broadcasting. 

Dr.  Carl  C.  Chambers,  Acting  Dean,  Moore  School  of  Electrical  Engineering,  Uni- 
versity of  Pennsylvania.  Mathematics,  Mathematical  Tables,  Units  and  Conversion  Fac- 
tors, Symbols  and  Abbreviations,  Constants,  Oscillators. 

R.  A.  Chegwidden,  Member  of  Technical  Staff,  BelHTelephone  Laboratories.  Magnetic 
Materials. 

H.  A.  Chinn,  Chief  Audio  Engineer,  Columbia  Broadcasting  System.  Audio  Facilities 
for  Sound  Systems,  Radio  Telephone  Broadcasting. 

E.  L.  Clark,  Sr.  Television  Engineer,  Radio  Corp.  of  America,  RCA  Victor  Division. 
Special  Purpose  Amplifiers. 

James  I.  Cornell,  Vickers  Electric  Division,  Vickers,  Inc.     Capacitors,  Condensers. 

L.  F.  Curtis,  Consultant,  Hazeltine  Electronics  Corp.     F-m  Receivers. 

B.  J.  Dalton,  Industrial  Engineering  Division,  Apparatus  Dept.,  General  Electric  Com- 
pany. Electronic  Control  Equipment. 

Paul  S.  Darnell,  Transmission  Apparatus  Engineer,  Bell  Telephone  Laboratories.  Re- 
sistors and  Rheostats. 

Howard  L.  Davis,  Jr.,  Engineer  in  Charge,  Special  Investigation  Section,  Engineering 
Department,  Philadelphia  Electric  Company.  Coordination  of  Communication  and  Power 
Systems. 

William  R.  Dohan,  General  Consultant,  Consulting  and  Standards  Division,  Engineer- 
ing Dept.,  Radio  Corp.  of  America,  RCA  Victor  Division.  Insulating  Materials. 

R.  D.  Duncan,  Jr.,  Engineering  Consultant.     F-m  Systems. 

Dr.  D.  W.  Epstein,  Research  Engineer,  Radio  Corp.  of  America,  RCA  Laboratories 
Division.  Geometrical  Optics,  Electron  Optics. 

J.  G.  Ferguson,  Engineer,  Bell  Telephone  Laboratories.  Measurement  of  Primary  Elec- 
trical Quantities. 

J.  L.  Finch,  Plant  Design  Superintendent,  Radio  Corp.  of  America,  RCA  Communica- 
tions, Inc.  Radio  Telegraph  Systems. 

H.  J.  Fisher,  Test  Engineer,  Bell  Telephone  Laboratories.     Wire  Line  Measurements. 

F.  J.  Gaffney,  Chief  Engineer,  Polytechnic  Research  and  Development  Co.,  Inc.    Signal 
Generators  and  Power  Measurements. 

Dr.  Maxwell  K.  Goldstein,  Air  Navigation  Development  Board,  CAA,  Dept.  of  Com- 
merce. Radio  Aids  to  Marine  Navigation. 

A.  J.  Grossman,  Member  of  Technical  Staff,  Bell  Telephone  Laboratories.  Electric 
Wave  Filters. 

0.  B.  Gunby,  Recording  Equipment  Systems  Engineer,  Radio  Corp.  of  America,  RCA 
Victor  Division.  Sound  Recording. 

A.  L.  Hammerschmidt,  Development  Engineer,  National  Broadcasting  Company,  Tele- 
vision Broadcasting. 

W.  B,  Hebenstreit,  Section  Head,  Navigation  and  Radar  Research,  Electronics  De- 
partment, Hughes  Aircraft  Company;  formerly  with  Bell  Telephone  Laboratories.  Mag- 


H.  A.  Henning,  Member  of  Technical  Staff,  Bell  Telephone  Laboratories.  Magnetic 
Recording  and  Reproducing  of  Sound,  Mechanical  Recording  and  Reproducing  of  Sound. 

L.  M.  Hershey,  Director  of  Research,  General  Instrument  Corporation.  Inductors  with 
Air  Cores. 


Vlll  LIST    OF   CONTRIBUTORS 

C.  J.  Hirsch,  Chief  Engineer,  Research  Division,  Hazeltine  Electronics  Corp.  /-/ 
Amplifier  a. 

E.  W.  Houghton,  Engineer,  Bell  Telephone  Laboratories.     Microwave  Measurements. 

Dr.  Herbert  E.  Ives,  Research  Consultant,  Bell  Telephone  Laboratories.  Photorespon- 
sive  Devices. 

A.  P.  Kauzmann,  Engineer,  Vacuum  Tube  Advanced  Development,  Radio  Corp.  of 
America,  RCA  Victor  Division.     Thermionic  Vacuum  Tubes. 

C.  R.  Keith,  Engineering  Representative,  ERP  Division,  Western  Electric  Company. 
Photographic  Sound  Recording. 

J.  P.  Kinzer,  Member  of  the  Technical  Staff,  Bell  Telephone  Laboratories.  Canty 
Resonators. 

Hugh  S.  Knowles,  Vice  President,  Jensen  Manufacturing  Company,  President  Indus- 
trial Research  Products,  Inc.  Effects  of  tfie  Acoustic  Medium,  Loudspeakers  and  Telephone 
Receivers,  Microphones. 

Vern  O.  Knudson,  Dean  and  Professor  of  Physics,  Graduate  Division,  University  of 
California.  Acoustic  Properties  of  Rooms,  Sound  Insulation,  Acoustic  Design  of  Auditoriums. 

R.  J.  Kowalski,  Commercial  Engineer,  Custom  Built  Dept.,  Radio  Corp.  of  America, 
RCA  Victor  Division.  Electro-acoustic  Equipment,  Public-address  Systems. 

V.  D.  Landon,  Radio  Research  Engineer,  Radio  Corp.  of  America,  RCA  Laboratories 
Division.  Single  Mesh  and  Coupled  Circuits,  Detectors,  Radio  Receivers. 

H.  W.  I/everenz,  Research  Chomico-Physicist,  Radio  Corp.  of  America,  RCA  Labora- 
tories Division.  Luminescent  and  Tenebrescent  Materials. 

B.  D.  Loughlin,  Sr.  Engineer,  Hazeltine  Electronics  Corp.    Distortion  and  Interference 
in  F-m  Systems. 

A.  V.  Loughren,  Vice  President  in  Charge  of  Research,  Hazeltine  Electronics  Corp. 
Television  Principles  and  Theory. 

Warren  A.  Harrison,  Member  of  Technical  Staff,  Bell  Telephone  Laboratories.  Fre- 
quency Measurements. 

Dr.  Warren  P.  Mason,  Research  Physicist,  Bell  Telephone  Laboratories.  Piezoelectric 
Crystals. 

Knox  Mcllwain,  Chief  Consulting  Engineer,  Hazeltine  Electronics  Corp.  Non-linear 
Electric  Circuits,  Electromagnetic  Radiation,  Electromechanical  Systems,  Conducting  Mate- 
rials, Single  Mesh  and  Coupled  Circuits. 

Henry  I.  Metz,  Chief,  Electronics  Maintenance  Branch,  Maintenance  Engineering  Div., 
Civil  Aeronautics  Administration.  Radio  Aids  to  Air  Navigation. 

H,  F.  Mickel,  Communication  Sales  Manager,  Raymond  Rosen  Engineering  Products, 
Inc.  Police  Radio. 

W.  A.  Munson,  Telephone  Engineer,  Bell  Telephone  Laboratories.  The  Sense  of  Hear- 
ing, Speech  and  Music,  Effect  of  Distortion. 

Dr.  Kenneth  N.  Ogle,  Optical  Research  Dept.,  Mayo  Clinic.     Vision. 

J.  J.  Okrent,  Sr.  Engineer,  Hazeltine  Electronics  Corp.     Pulse  Circuits. 

D.  S.  Peck,  Member  Technical  Staff,  Bell  Telephone  Laboratories.    Gaseous  Conduc- 
tion Tubes. 

Harold  Pender,  Consultant,  Moore  School  of  Electrical  Engineering,  University  of 
Pennsylvania. 

Dr.  E.  Peterson,  Member  Technical  Staff,  Bell  Telephone  Laboratories.    Wave  Analysis. 

J.  D.  Phyfe,  Product  Engineer,  Theatre  Sound  Equipment,  Radio  Corp.  of  America, 
RCA  Victor  Division.  Projection  Practices. 

N.  Y.  Priessman,  Member  of  Technical  Staff,  Bell  Telephone  Laboratories.  Varistors 
and  Thermistors. 

Dr.  G.  L.  Ragan,  Research  Associate,  General  Electric  Company.  Wave-guide  Compo- 
nents. 

Dr.  E.  G.  Ramberg,  Research  Physicist,  Radio  Corp.  of  America,  RCA  Laboratories 
Division.  Television  Pick-up  Tubes. 

P.  H.  Richardson,  Member  of  Technical  Staff,  Bell  Telephone  Laboratories.  Theory  of 
Linear  Passive  Networks,  Recurrent  Networks. 

Arnold  J.  Rohner,  Project  Engineer,  Bendix  Radio  Division.  Ferrous  Cored  Inductors, 
Transformers  with  Iron  Cores. 

A.  L.  Samuel,  Professor  of  Electrical  Engineering,  University  of  Illinois.     Klystrons. 

Arthur  H.  Schafer,  Member  of  Technical  Staff,  Bell  Telephone  Laboratories.  Resistors 
and  Rheostats. 

Dr.  S.  A.  Schelkunoff,  Member  of  Technical  Staff,  Bell  Telephone  Laboratories.  Wave- 
guide Theory. 

J.  C.  Schelleng,  Radio  Research  Engineer,  Bell  Telephone  Laboratories.  Radio  An- 
tennas, Transmission  in  Space. 


LIST    OF    CONTRIBUTORS  IX 

Dr.  Theodore  Seller,  Professor  of  Physics,  Amherst  College.    Cathode-ray  Tube  Displays, 

F.  J.  Somers,  Staff  Engineer,  National  Broadcasting  Company.    Television  Broadcasting* 

Dr.  John  C.  Steinberg,  Member  of  Technical  Staff,  Bell  Telephone  Laboratories.  The 
Sense  of  Hearing,  Speech  and  Music,  Effects  of  Distortion. 

L.  E.  Swedlund,  Sr.  Engineer,  Radio  Corp.  of  America,  RCA  Victor  Division.  Cathode- 
ray  Tubes. 

W.  O.  Swinyard,  Chief  Engineer,  Hazeltine  Research,  Inc.  Routine  Measurement  of 
A-m  and  F-m  Broadcast  Receivers. 

John  D.  Taylor,  Consulting  Engineer,  American  Telephone  &  Telegraph  Company. 
Telephony,  Telegraphy,  Coordination  of  Communication  and  Power  Systems,  Wire  Trans- 
mission  Lines,  Mechanical  Features  of  Transmission  Lines. 

L.  Vieth,  Member  of  Technical  Staff,  Bell  Telephone  Laboratories.  Magnetic  Recording 
and  Reproducing  of  Sound,  Mechanical  Recording  and  Reproducing  of  Sound. 

Arthur  H.  Volz,  Member  of  Technical  Staff,  Bell  Telephone  Laboratories.  Potentiom- 
eters and  Rheostats. 

Dr.  S.  Reid  Warren,  Jr.,  Associate  Professor,  Moore  School  of  Electrical  Engineering, 
Associate  Professor  Radiologic  Physics,  Graduate  School  of  Medicine,  University  of  Penn- 
sylvania. X-ray  Tubes,  Medical  Applications  of  Electricity. 

Charles  Weyl,  Professor,  Moore  School  of  Electrical  Engineering,  Associate  Professor 
of  Radiologic  Physics,  Graduate  School  of  Medicine,  University  of  Pennsylvania.  Med- 
ical Applications  of  Electricity. 

H.  A.  Wheeler,  Consulting  Radio  Physicist,  engaged  for  this  work  by  Hazeltine  Elec- 
tronics Corp.  Pulse  Techniques,  Transient  Networks. 

I.  G.  Wilson,  Member  of  Technical  Staff,  Bell  Telephone  Laboratories.  Cavity  Reso- 
nators. 

J.  E.  Young,  Supervisor,  Broadcast  Engineering  Section,  Radio  Corp.  of  America,  RCA 
Victor  Division.  Modulators,  Power  Supply,  Radio  Transmitters,  F-m  Transmitters. 

Dr.  V.  K.  Zworykin,  Vice  President  and  Technical  Consultant,  Radio  Corp.  of  America, 
RCA  Laboratories  Division.  Television  Pick-up^Tubes. 


GENERAL  TABLE  OF  CONTENTS 


Detailed  tables  of  contents  are  given  at  the  beginning  of  each  section.     An  alphabetical  index  appears 
after  Section  SS. 


SECTION  1.     MATHEMATICS,  UNITS, 
AND  SYMBOLS 

PAGE 

Mathematics , 1-02 

Mathematical  Tables  and  Charts 1-19 

Units  and  Conversion  Factors 1-42 

Symbols  and  Abbreviations 1-71 

Constants 1-79 

SECTION  2.    PROPERTIES  OF  MATERIALS 


Conducting  Materials . 
Insulating  Materials . . 
Magnetic  Materials . , , 


2-02 
2-21 
2-56 


SECTION  3.    RESISTORS,  INDUCTORS, 
AND  CAPACITORS 


Resistors  and  Rheostats . . . 
Varistors  and  Thermistors. 
Inductors  with  Air  Cores .  . 
Ferrous-cored  Inductors . .  . 
Capacitors 


3-02 
3-22 
3-31 
3-42 
3-53 


SECTION  4.  ELECTRON  TUBES 


Thermionic  Vacuum  Tubes . 

Magnetrons 

Klystrons 

Gaseous  Conduction  Tubes . 
X-ray  Tubes 


4-02 
4-40 
4-51 
4-58 
4-81 


SECTION  6.     ELECTRIC  CIRCUITS, 
LINES,  AND  FIELDS 

Theory  of  Linear  Passive  Networks ....         5-02 

Recurrent  Networks 5-22 

Transients  in  Networks 5-26 

Non-linear  Electric  Circuits 5-37 

Electromagnetic  Radiation 5-49 

Electromechanical  Systems 5-56 

SECTION  6.    PASSIVE  CIRCUIT  ELEMENTS 

Single-mesh  and  Coupled  Circuits 6-02 

Transformers  with  Iron  Cores 6-13 

Electric  Wave  Filters 6-33 

Badio  Antennas 6-62 

SECTION  7.    VACUUM-TUBE  CIRCUIT 
ELEMENTS 

Amplifiers 7-02 

Special-purpose  Amplifiers 7-31 

Intermediate-frequency  (I-F)  Amplifiers         7-56 

Modulators 7-70 

Detectors 7-76 

Oscillators 7-83 

Power  Supply 7-106 

Radio  Receivers 7-115 

Radio  Transmitters 7-128 

SECTION   8.     FREQUENCY   MODULATION 

Frequency-modulation  Systems 8-02 

Frequency-modulation  Transmitters 8-09 

Frequency-modulation  Receivers 8-16 

Distortion  and  Interference  in  F-M  Sys- 
tems           8-26 


SECTION  9.    PULSE  TECHNIQUES 


Pulses  and  Pulse  Systems. 
Pulse  Circuits '. 


PAGE 

9-02 
9-13 


SECTION  10.    TRANSMISSION  CIRCUITS 

Wire  Transmission  Lines 10-02 

Wave  Guides— Theory 10-09 

Wave-guide  Components 10-17 

Transmission  in  Space 10-29 

Mechanical  Features  of  Transmission 

Lines 10-49 

Coordination  of  Communication  and 

Power  Systems 10-67, 

SECTION  11.    ELECTRICAL 
MEASUREMENTS 

Frequency  Measurements 11-02, 

Measurement  of  Primary  Electrical 

Quantities 11-16 

Wire  Line  Measurement 11-32 

Routine  Measurements  on  A-M  and  F-M 

Broadcast  Receivers 11-43 

Wave  Analysis 11-54 

Microwave  Measurements 11-69 

Signal  Generators  and  Power  Measure- 
ment   11-89 

SECTION  12.     ACOUSTICS 

The  Sense  of  Hearing 12-02 

Speech  and  Muaic 12-19 

Effects  of  Distortion  on  Speech  and  Music  12-29 

Acoustic  Properties  of  Rooms 12-39 

Sound  Insulation 12-57 

Acoustic  Design  of  Auditoriums 12-69 

SECTION  13.     ELECTROMECHANICAL- 
ACOUSTIC  DEVICES 

Effects  of  the  Acoustic  Medium 13-02 

Loudspeakers  and  Telephone  Receivers .  13-08 

Microphones 13-22 

Magnetic  Recording  and  Reproducing  of 

Sound 13-28 

Mechanical  Recording  and  Reproducing 

of  Sound 13-37 

Photographic  Sound  Recording 13-47 

Piezoelectric  Crystals 13-55 

SECTION  14.     OPTICS 

Geometrical  Optics 14-02 

Vision 14-25 

Electron  Optics 14-49 

SECTION  15.     ELECTRO-OPTICAL 
DEVICES 

Photoresponsive  Devices 15-02 

Television  Pick-up  Tubes 15-19 

Luminescent  and  Tenebrescent  Materials  15-29 

Cathode-ray  Tubes 15-41 


Xil 


GENERAL   TABLE    OF   CONTENTS 


SECTION  16.     SOUND-REPRODUCTION 
SYSTEMS 

PAGE 

Audio  Facilities  for  Sound  Systems 16-02 

Electroacoustic  Equipment 16-11 

Public-address  Systems 16-14 

Sound  Recording  and  Projection 16-19 

Radio  Telephone  Broadcasting 16-25 

Police  Radio 16-35 

SECTION  17.     TELEPHONY 

Central-office  Equipment 17-03 

Radio  Telephone  Systems 17-54 

Telephone  Lines — Transmission  Consid- 
erations    17-69 

Program  Service 17-101 

Subscriber  Stations 17-106 

SECTION  18.    TELEGRAPHY 

Theory 18-02 

Telegraph  Systems 18-18 

Submarine  Cable  Telegraphy 18-40 

Telegraph  Equipment 18-46 

Transmission  Maintenance 18-53 

Radio  Telegraph  Systems 18-56 

SECTION  19.     FACSIMILE  TRANSMISSION 
AND  RECEPTION 

Scanning  Systems 19-02 

Recording  Systems 19-11 

Synchronizing  and  Phasing 19-18 

Transmission  Characteristics 19-22 

Specialized  Applications 19-23 


SECTION  20.     TELEVISION 

PAGE 

Principles  and  Theory 20-02 

Television  Broadcasting 20-21 

Television  Receivers 20-46 

Other  Forms  of  Television 20-64 


SECTION  21.     ELECTRONIC  CONTROL 
EQUIPMENT 

Fundamental  Electronic  Power  Circuits  21-02 
Fundamental  Electronic  Control  Circuits  21-13 
Complete  Electronic  Devices 21-20 


SECTION  22.     AIDS  TO  NAVIGATION 

Radio  Aids  to  Air  Navigation 22-04 

Radio  Aids  to  Marine  Navigation 22-33 


SECTION  23.     MEDICAL  APPLICATIONS 
OF  ELECTRICITY 

Electrotherapy  and  Shock  Therapy ....  23-02 
Diathermy  and  High-frequency  Surgery  23-04 
The  Medical  Uses  of  Ultraviolet  and  In- 
frared Radiations 23-06 

Electrocardiography  and  Electroenceph- 

alography 23-08 

Electroacoustic  Devices 23-11 

Roentgen  Therapy 23-12 

Roentgenography  and  Roentgenoscopy .  23-14 

High- voltage  Shock  and  X-ray  Burn .  . .  23-17 


Tills  book  Is  divided  into  sections,  each  section 
carrying  its  in. dependent  seqnen.ce  of  page  numbers. 
For  example,  3— 315  in.dica.tes  Section  3,  page  ±5. 


SECTION  1 
MATHEMATICS,  UNITS,  AND  SYMBOLS 


MATHEMATICS 
ART.  BY  CARL  C.  CHAMBERS  PAGE 

1.  Algebraic  Formulas 02 

2.  Complex  Quantities 06 

3.  Trigonometric  Formulas 07 

4.  Exponential  and  Hyperbolic  Formulas. .  10 

5.  Calculus  Formulas 12 

6.  Differential  Equations 13 

7.  Errors  of  Observation 15 

8.  Approximations 16 

9*  Series 17 

10.  Mensuration 17 

MATHEMATICAL  TABLES  AND  CHARTS 

11.  Common    and    Natural    Logarithms    of 

Numbers 19 

12.  Trigonometric  Tables 21 

13.  Exponential  and  Hyperbolic  Tables ....  26 

14.  Bessel  Functions 37 

15.  Transmission  Unit  and  Power  Reference 

Levels. — Decibels 37 

UNITS  AND   CONVERSION  FACTORS 

BY  J.  G.  BRAINERD  AND  CARL  C.  CHAMBERS 

16.  Systems  of  Units 42 

17.  Conversion  Tables 47 

Table 

1.  Length  (L) 47 

2.  Area  (L2) 48 

3.  Volume  (L3) 49 

4.  Plane  Angle  (No  Dimensions') . .  51 

5.  Solid  Angle  (No  Dimensions) .  .  51 

6.  Time  (T) 51 

7.  Linear  Velocity  (LT~l) 52 

8.  Angular  Velocity  (T~l) 53 

9.  Linear  Acceleration  (LT~2) ...  53 

10.  Angular  Acceleration  (T7""2) 53 

11.  Mass  (M)  and  Weight 54 

12.  Density  or  Mass  per  Unit  Vol- 

ume (,MX~3) 55 

13.  Force  (MLT~2)  or  (F) 55 

14.  Torque    or    Moment    of   Force 

(ML2T~2)  or  (FL) 56 

15.  Pressure  or  Force  per  Unit  Area 

(ML-1T~2)  or  (FL~2) 56 

16.  Energy,    Work    and    Heat 

(ML2T~2)  or  (FL) 57 

17.  Power  or  Rate  of  Doing  Work 

(ML2T~*)  or  (FLT~l) 58 

18.  Quantity     of    Electricity     and 

Electric  Flux  (Q) 58 


Table 
19. 

20. 
21. 
22. 

23. 

24. 
25. 
26. 

27. 

28. 
29. 
30. 
31. 
32. 


35. 
18.  Gages. 


Charge    per     Unit    Area     and 

Electric  Flux  Density  (QI/~2)     59 
Electric  Current  (QT~l)  .......      59 

Current  Density  (QT~1L~2)  ,  .  .      59 
Electric  Potential  and  Electro- 
motive Force   (MQ~1L2T~2) 
~ 


Electric  Field  Intensity  and  Po- 
tential Gradient  (MQ^LT'2) 
or  (FQ-1)  ................. 

Electric  Resistance 
(MQ~2L2T~^  or  (FQ~2LT)  . 

Electric  Resistivity 
(MQ~WT~l)  or  (FQ~2L2T) 

Electric    Conductivity 
(M-1Q2L~3T) 
(F~1Q2L-* 

Capacitance 
-- 


Inductance    (MQ~2L2)    or 
(FQ~2LT2)  ................ 

Magnetic  Flux  (MQ^L2^1}  or 


Magnetic  Flux  Density 
(MQ-iT-1)  or  (FQ~1L~1T) . 

Magnetic  Potential  and  Mag- 
netomotive Force  (QT~l) 

Magnetic  Field  Intensity,  Po- 
tential Gradient,  andJVtagne- 
tizing  Force  (QL~~1T~1') 

Specific  Heat  (L2T~2t~ 


60 

60 
61 
61 

62 
62 
63 
63 
63 
64 


64 
64 

Thermal  Conductivity 
(MLT~zrl)  and  Thermal  Re- 
sistivity (Af"~1zr~1:r3i)  (t  = 

Temperature) 65 

Light 65 

66 


SYMBOLS  AND  ABBREVIATIONS 

19.  Abbreviations  for  Engineering  Terms. . .  71 

20.  Letter  Symbols  for  the  Magnitudes   of 

Electrical  Quantities 72 

21.  Standard  Graphical  Symbols 76 

22.  Use  of  Greek  Alphabet  for  Symbols 79 

CONSTANTS 

BY  CARE,  C.  CHAMBERS 

23.  Principal  Physical  Constants  and  Ratios     79 

24.  Standard  Radio-frequency  Ranges 80 


1-01 


MATHEMATICS,  UNITS,  AND   SYMBOLS 
MATHEMATICS 

By  Carl  C.  Chambers 

1.    ALGEBRAIC  FORMULAS 

MISCELLANEOUS   FORMULAS 

(a  db&)2  -  a2dh2a&  +  62 

(a  d=  6)3  =  a3  =b  3a2  6  +  3a62  db  &3 

(a  =b  b)n  -2   feifrTife)!  a*  (±rfn~®*        n!  «  »(n  -  1)  .  .  .  3  X  2  X  1 

a2  -  62  «  (a-f  6)  (a-  6)  _ 

a2  +  52  =  (a  -f  y&)(0  -y&),    y  «  V-  i 
a31  X  ay  =  ate4v)f     a°  =  1  [for  a  5*  0),     (a&)*  =  a*  6* 


-  _ 

a"  '  "a"     \b 

a1/* 


log  (a31)  =  a;  log  a,     log  #&  =  log  a  +  log  & 
log  7  ^  log  a  —  log  6 

r£  a       c  a±6       c±d         .a  — 

If  7  ~  -    then     —  r  —  =  —  —     and 


~  —  r  —      —  —  —  r—  -  =  —  ;  —  _ 

b      d  b  d  a+b       c  +  d 

The  sum  of  an  arithmetical  progression  is  given  by 

*  =  £(<*  +  *)  -|{2a+(n-  !)<*} 

whore  Z  —  a  +  (n  —  l)d  is  the  last  term,  a  is  the  first  term,  d  is  the  common  difference, 
and  «  is  tho  sum  of  the  n  terms. 

The  sum  of  a  geometrical  progression  is  given  by 

(1  -  rn)       Ir  -  a 

s  —  a  —  -  =  -  - 

1  —  r          r  —  1 

where  Z  »  arn-1  is  the  last  term,  a  is  the  first  term,  d  is  the  common  ratio,  and  s  is  the  sum 
of  the  n  terms.     If  n  approaches  infinity  and  r2  is  less  than  unity 


The  multiple  product  represented  by  n(n  —  l)(n  —  2)  .  .  .  3  X  2  X  1  is  designated 
by  the  symbol  n!  or  |n  and  is  called  "  n  factorial."     The  following  list  gives  the  value  of 

n!  up  to  n  =  10 

1!  -      1  6!  =  720 

2!  =       2  7!  =  5,040 

3!  =       6  8!  =  40,320 

41  =24  9!  «  362,880 

5!  -  120  10!  =  3,628,800 
1-02 


ALGEBRAIC   FORMULAS  1-03 

For  large  values  of  n  a  good  approximation  for  nl  is,  from  Stirling's  formula, 
n\  =  (2*7i)  **  (-Y,     e  =  2.7182818 

This  formula  is  accurate  to  about  21f2  per  cent  at  n  =  10  and  becomes  more  accurate 
very  rapidly  as  n  is  increased, 

The  number  of  permutations  or  arrangements  of  n  things  taken  p  at  a  time  is 


pn 
* 


The  number  of  combinations  of  n  things  taken  p  at  a  time  is  then 


QUADRATIC  EQUATION.     The  solution  of 

ax2  +  bx  +  c  =  0  

—6  db  V&2  -  4ac 
is  re  = 

If  a,  &,  and  c  are  real,  and  the  discriminant,  62  —  4ac,  is  positive,  the  roots  are  real  and 
unequal;  if  it  is  zero,  the  roots  are  real  and  equal;  if  it  is  negative,  the  roots  are  conjugate 
complex  numbers. 

CUBIC  EQUATIONS.     The  solution  of 

is  obtained  as  follows:"  Put  x  =  -  (y  —  6) ;  then  (1)  becomes 

a 

y3  -  3Hy  +  0  =  0 
where  H  =  62  —  ac 

G  =  a2  d  -  3abc  +  2bs 
For  a  solution  let 

/? 

then  the  values  of  y  will  be  given  by 

1  Vs  1  Vs 

y  =  A  +  B.     —  -  (A  +  B)  +  j  -r-  (A  —  £),     —  -  (A  +  B)  —  j (A  —   J?) 

2222 

If  a,  b,  c,  d  are  real  and  if  (72  —  4F3,  the  discriminant,  is  positive  there  are  one  real 
root  and  two  conjugate  complex  roots;  if  G2  —  4#3  is  zero  there  are  three  real  roots,  at 
least  two  of  which  are  equal;  if  (r2  —  4#3  is  negative  there  are  three  real  and  unequal 
roots. 

The  solution  may  be  written  in  three  other  forms. 

(1)  Put 

*-5«- 
then  the  roots  are 


y  =  2  VSsin  0,     2  V^T  sin  (0  +  120°),     2  v^sin  (0  -  120°) 
Or  (2)  put 

le^rai 

3  L2#v#J 

then  the  roots  are 

y  =  —  2  V]j  cosh  w,     ^~H  cosh  w  +  V— 3H  sinh  u,     Vj?  cosh  u  —  V—  3#  sinh  u 

Or  (3)  put 

1   .  ,.r G 

u  =  -  smh  1 1  — 


Then  the  roots  are 

^  =  2  V— H  sinh  w,     —  V— #  sinh  w  +  V§F  cosh  w 

—  V— ^  sinh  w  —  VsH  cosh  w 

SIMULTANEOUS  EQUATIONS.     Given  7^  independent  equations  in  n  unknowns, 
these  n  equations  usually  fix  one  or  more  values  for  each  of  the  n  unknowns.     To  solve 


1-04 


MATHEMATICS,   UNITS,   AND   SYMBOLS 


such,  a  set  of  simultaneous  equations  in  x,  y,  and  2,  say,  solve  each  of  the  three  equations 
for  x  in  terms  of  y  and  z.  Equating  these  three  values  for  x  gives  two  equations  in  y 
and  z.  Solving  each  of  these  two  equations  for  y  in  terms  of  z  and  equating  these  two  values 
of  y  gives  a  single  equation  in  z.  The  solution  of  this  last  equation  then  gives  the  value 
of  z.  Then  substitute  this  value  of  z  in  either  of  the  equations  in  y  and  z,  and  solve  for  y. 
Then  substitute  these  values  of  y  and  z  in  any  one  of  the  original  equations  and  solve  for  x. 

DETERMINANTS.  In  the  case  of  linear  simultaneous  equations  (i.e.,  when  x,  y, 
and  z  occur  only  to  the  first  power),  the  equations  may  be  solved  by  determinants.  This 
method  is  a  considerable  time-saver  when  the  number  of  unknowns  is  greater  than  three, 
but  when  the  number  of  unknowns  is  three  or  less  the  straight  substitution  method  is 
preferable. 

The  determinant  of  a  set  of  simultaneous  equations  is  formed  by  writing  the  equations 
one  below  the  other  with  the  same  unknown  in  the  same  relative  position  in  each.  The 
block  of  numbers  forming  the  coefficients  of  the  unknowns  is  called  the  determinant. 
For  example,  the  determinant  of  the  equations 


w  + 
to  "h 
w  -f 
w  4- 


D 


2/+    2=6 

y  -f"  3z  =  4 

3y  =  1 

9  =  3 


1111 
1013 
1230 
1301 


The  values  of  any  one  of  the  unknowns,  say  y,  is  found  by  writing  a  second  determinant, 
Dy,  exactly  like  the  determinant  D,  except  that  the  constants  forming  the  right-hand 
members  of  these  equations  are  substituted  for  the  coefficients  of  y  in  the  determinant, 
that  is 

1     6 


Then 


and  similarly  for  the  other  unknowns. 

The  value  of  any  determinant  is  found  by  making  use  of  the  following  rules: 

(1)  If  a  determinant  has  two  equal  rows  or  columns,  it  is  equal  to  zero. 

(2)  To  any  row  or  column  one  may  add  or  subtract  any  number  of  times  any  other 
row  or  column  without  altering  the  value  of  the  determinant. 

(3)  To  multiply  any  row  or  column  by  a  number  is  the  same  as  multiplying  the 
determinant  by  that  number. 

(4)  If  all  the  terms  in  a  row  or  column  except  one  are  zero,  the  determinant  reduces 
to  one  of  a  lower  order  which  may  be  obtained  by  striking  out  the  row  and  column  which 
intersect  at  the  element  of  the  row  or  column  which  is  not  zero,  and  multiplying  the  whole 
by  that  element,  changing  the  sign  of  this  element,  however,  if  it  is  removed  by  an  odd 
number  of  elements  from  the  principal  diagonal.     The  principal  diagonal  is  the  line  of 
elements  beginning  at  the  upper  left-hand  corner  and  ending  at  the  lower  right-hand 
corner.     Thus, 


the  principal  diagonal  being  that  with  the  figures  1,4,  0,  and  3.     It  is  immaterial  whether 
the-  distance  from  the  diagonal  is  counted  along  a  row  or  a  column. 
(5)  The  value  of  a  determinant  of  the  second  order  is 

0,2       63 

The  reduction  of  determinants  is  effected  by  altering  the  terms  according  to  the  above 
rules  until  a  row  or  column  is  obtained  in  which  all  terms  but  one  are  zero.  This  enables 
a  reduction  of  order  to  be  effected  in  accordance  with  rule  4.  Reductions  are  continued 
until  one  of  the  second  order  is  obtained. 


ALGEBRAIC  FORMULAS 
EQUATIONS  OF  COMMON  CURVES.    Straight  Line. 


y  =  x  tan  6  -f-  6. 


Circle. 


Ellipse. 


a2      &2 


Parabola  (Vertical). 

y-  fcz2 
where  k  is  a  constant. 

Parabola  (Horizontal). 

y  «  &V^ 
where  &  is  a  constant. 

Hyperbola. 

a2       2/2 

"5  "~  75  =  1  (Horizontal) 
6        ^ 


Rectangular  or  Equilateral  Hyperbola. 


where  ^  is  a  constant. 


Catenary. 


r  cosh  fca;  — 


where  /c  is  a  constant.     The  length  of  arc 
from  O  to  P  is 

«=»  y  sinh  (kx) 
fc 

See  tables  of  hyperbolic  functions. 


-rX 


Sinusoid. 


^Lsin  (ax  +  0). 


,1-05 


1-06  MATHEMATICS,   UNITS,    AND   SYMBOLS 

2.    COMPLEX  QUANTITIES 

The  square  root  of  a  negative  quantity  is  called  an  "  imaginary  "  quantity,  or  a  pure 
imaginary.  A  quantity  consisting  of  the  sum  or  difference  of  a  real  quantity  and  an 
imaginary  quantity  is  called  a  "  complex  "  quantity.  All  the  rules  of  ordinary  algebra 
apply  to  pure  imaginaries  and  complex  quantities.  The  square  root  of  minus  one  is 
called  the  imaginary  unit  and  is  usually  represented  by  the  symbol  j  (writers  on  pure 
mathematics  use  the  symbol  i),  that  is, 


Any  complex  quantity  may  then  be  written 

a+jb 
where  a  and  6  are  both  real  quantities. 

GEOMETRICAL  REPRESENTATION  OF  A  COMPLEX  QUANTITY.  A  positive 
real  quantity  may  be  represented  by  a  line  drawn  on  a  plane  in  a  given  direction;  a  negative 
real  quantity  may  be  represented  by  a  line  drawn  in  the  opposite  direction.  Multiplying 
a  quantity  by  —  1  then  reverses  its  direction.  Also,  since  multiplying  a  real  quantity  by 
v'— 1  twice  is  equivalent  to  multiplying  it  by  —  1,  the  operation  of  multiplying  once  by 
v— 1  may  be  represented  by  turning  the  line  representing  the  quantity  through  90° 
in  the  positive  direction  of  rotation.  The  positive  direction  of  rotation  is  taken  as  the 
opposite  direction  to  that  in  which  the  hands  of  a  clock  move.  Hence,  a 
complex  quantity  a  +  jb  may  be  represented  by  the  line  OP  in  the 
figure,  where  OA  =  a  and  AP  =  b.  The  complex  quantity  a  +  jb  is 
then  completely  specified  by  a  line  of  length  v  a2  +  bz  making  an  angle 

6,   with    the    axis    of    reference    OX    where    tan  0  =  -,    The  length 

a 

f  =  Va2 -{•- 62   is    called    the    magnitude   of   the    complex    quantity,    and    the    angle 

0  =  tan""1  -  is  called  its  angle.     From  the  figure  it  is  evident  that  the  complex  quantity 
a 

a  +  jb  may  also  be  written 

a  +  jb  —  M  (cos  6  +j  sin  0) 

Expanding  cow  0  and  sin  0  into  vSeries  (see  Series,  Article  9)  and  adding,  the  resultant  series 
obtained  is  the  series  for  e'e ;  hence 

a  +  jb  -  Me1'9  (1) 

From  the  above  definitions  and  equation  (1)  it  is  evident  that  complex  numbers 
possess  the  following  properties: 

ADDITION   OF   TWO    COMPLEX   QUANTITIES. 

(a  -f-  jb)  +  (ai  +  jbi)  =  (a  +  ai)  +  /(&  +  &i) 
SUBTRACTION   OF  TWO    COMPLEX   QUANTITIES. 

(a  +  jb)  —  (ai  +  jbi)  =  (a  —  ai)  +  j(b  —  61) 
MULTIPLICATION  OF  A  COMPLEX  QUANTITY  BY  A  COMPLEX  NUMBER. 

(a  +  jb)  (ai  +  jbi)  —  aai  —  bbi  +  j(abi  +  a\  6) 
or,  putting  a  +  ft  =  Me1'9     and    en  +  jbi  =  Mi  e^1 

whore  M  = 


and  tan  0i  =  — 

ai 

we  have  (a  +  jb)  (01  +  ybi)  =  M49  Ml  </dl  -  Jlf  MI 

Honco  the  product  of  two  complex  quantities  is  in  general  a  complex  quantity  which  has 
a  magnitude  equal  to  the  product  of  the  magnitudes  of  the  two  quantities  and  an  angle 
equal  to  tho  sum  of  the  angles  of  the  two  quantities. 

DIVISION   OF  A   COMPLEX    QUANTITY  BY  A   COMPLEX  NUMBER. 

a  +  Jb    _    (a  +  ;'?>)  (ai  —  y&i)    _  aai  +  bbi  —  y(abi  —  ai  b) 
""  i)  (ai  —  fti)  ~"  ^  +  &i2 

M3Q         M_  j(e- 6i) 
Mi<Jdl==S  Mi 


TEIGONOMETRIC  FORMULAS 


1-07 


Hence  the  quotient  of  two  complex  quantities  is  in  general  a  complex  quantity  which  has  a 
magnitude  equal  to  the  quotient  of  the  magnitudes  of  the  two  quantities  and  an  angle 
equal  to  the  difference  of  the  angles  of  the  two  quantities. 
SQUARE  ROOT  OF  A  COMPLEX   QUANTITY. 


and  in  general 


.     /Va2  +  62  -  a] 

'V — i — J 


a2  +  fr2  +  a        .     /Va2  +  b2  -~a] 
—2 >V 2 J 


Hence  the  nth  root  of  a  complex  quantity  is,  in  general,  a  complex  quantity  which  has 
a  magnitude  equal  to  the  nth  root  of  the  magnitude  M  of  the  quantity  and  an  angle  equal 
one-nth  of  the  angle  of  the  quantity. 

EQUATIONS  CONTAINING  COMPLEX  QUANTITIES.  Since  a  real  quantity  can- 
not be  equal  to  an  imaginary  quantity  it  follows  that  any  equation  of  the  form 

A  +  JB  -  Ai  +  jBi 

where  A,  B,  A\,  and  Bi  are  all  real  quantities  (which  may,  however,  consist  of  any  number 
of  terms)  ,  is  equivalent  to  the  two  equations 

A  =  Ai 
and 


Also,  if  one  member  of  an  equation  reduces  to  the  form  A  -f-  jB,  then  the  other  member  of 
this  equation  must  likewise  contain  an  equal  real  and  an  equal  imaginary  part. 

See  K.  S.  Johnson,  Transmission  Circuits  for  Telephonic  Communication,  D.  Van  Nos- 
trand. 

3.  TRIGONOMETRIC  FORMULAS 

The  trigonometric  functions  of  an  angle  are  the  ratios  to  one  another  of  the  various 
sides  of  a  right  triangle  having  the  given  angle  as  one  of  its  angles.  Kef  erring  to  Fig.  1, 
let  B,  P,  and  H  be  the  three  sides  of  a  triangle.  Then  the  trigonometric  functions  of  the 
angle  x  are 


sine  of  x,  abbreviated     -    sin  x  =  ~ 


cosine  of  x,  abbreviated     cos  x  =  — 


tangent  of  re,  abbreviated  tan  x  —  — 


cotangent  of  x,  abbreviated  cot  x  =  — 


secant  of  x,  abbreviated        sec  x  =  — 


cosecant  of  x,  abbreviated    esc  x  =  — 


When  B,  P,  and  H  are  limited  to  the  three  sides  of  a  right  triangle,  the  above  definitions 
are  directly  applicable  only  to  angles  lying  between  0  and  90°.  The  definitions,  however, 
may  be  extended  by  considering  the  point  A  (Fig.  2)  as  describing  a  circle  of  radius  OA 


B    „ 
FIG.  1 


FIG.  2 


with  the  center  at  0.    Let  XX'  be  the  horizontal  diameter  and  YYf  the  vertical  diameter 
of  this  circle,  and  call  P  the  perpendicular  distance  from  A  to  the  line  XX'  and  B  the 


1-08 


MATHEMATICS,   UNITS,  AND   SYMBOLS 


horizontal  distance  from  A  to  YY'.  P  is  to  be  considered  positive  when  A  lies  above 
XX',  negative  when  below.  B  is  considered  positive  when  ,4  is  to  the  right  of  FF7  and 
negative  when  to  the  left.  The  four  quarters  of  the  circle  are  called  quadrants,  and  are 
designated  as  the  first,  second,  third,  and  fourth  quadrants  as  indicated.  The  angle  is 
said  to  lie  in  the  quadrant  in  which  the  point  A  lies.  In  Fig.  2  the  angle  x  is  in  the  second 
quadrant. 

Algebraic  Signs  of  the  Functions 


Sine 

Cosine 

Tangent 

Angle  in  first  quadrant   

+ 

+ 

+ 

Angle  in  second  quadrant  

+ 

Angle  in  third  quadrant  

_ 

+ 

Angle  in  fourth  quadrant  

- 

+ 

Period.  From  the  above  definitions  it  is  evident  that  adding  2?r  radians  or  360°  to  an 
angle  does  not  change  the  value  of  any  of  its  functions;  that  is,  these  functions  repeat 
themselves  every  time  the  angle  increases  by  the  2ir  radians  or  360°.  They  are  therefore 
said  to  have  a  period  equal  to  2ir  radians  or  360°. 

Functions  of  Angles  in  Any  Quadrant  in  Terms  of  Angles  in  First  Quadrant. 


sin  (90  +  #)  =  cos  x 

cos  (90  +  re)  =  —  sin  x 

tan  (90  +  x)  =  -  cot  x 

sin  (180  4-  re)  -  —  sin  x 

cos  (180  -f  x]  =  —coax 

tan  (180  +  x)  =  tan  x 

sin  (270  +  a;)  ==  —  cos  x 

cos  (270  -f-  re)  =  sin  x 

tan  (270  +  x]  =  -  cot  x 


sin  ( —  x)  —  —  sin  x 
cos  ( —  re)  —  cos  x 
tan  ( —  x]  =  —  tan  x 
sin  (180  —  x)  —  sin  x 
cos  (ISO  —  x]  —  —  cos  x 
tan  (180  -  x)  =  -  tan  x 
sin  (270  —  x)  =  —  cos  x 
cos  (270  -  x)  -  —  sin  x 
tan  (270  -  x)  =  cot  x 

Anti-functions.  If  a  =  sin  x,  then  x  is  the  angle  whose  sine  is  a;  this  may  be  expressed 
symbolically  x  =  sin"1  a,  which  is  read  "x  equals  the  angle  whose  sine  is  a."  The  angle  x 
is  also  called  the  "anti-sine"  or  the  "inverse  sine"  of  a.  Similar  notation  is  used  for  the 
other  functions;  for  example,  x  —  cos™1  b  is  used  to  express  the  relation  that  x  is  the  angle 
whose  cosine  is  b.  At  least  two  "anti-functions"  must  be  known  to  completely  determine 
the  quadrant  in  which  an  angle  lies;  for  example,  if  x  —  sin""1  0.5  then  x  may  be  either  30° 
or  150°,  but  if  we  also  have  x  =  cos"1  0.866,  then  x  must  equal  30°,  while  if  x  -  cos"1 
(-0.866),  then  re  must  equal  150°. 

Anti-functions  may  be  taken  from  the  Trigonometric  Tables  by  finding  the  angle  in  the 
margin  corresponding  to  the  function  in  the  table. 

Example,     sin"1  0.319  «  18.6°  or  180°  -  18.6°  =  161.4°. 

Versine.     The  expression  (1  —  cos  x)  is  called  the  "versine"  of  x. 

Relations  among  Functions  of  the  Same  Angle. 


x              sin  x 

1 

cos  re 

-                     sin  re  4~  cos  re  — 
cot  x 

1 

11       +01-»2    n. 

1 

sec  re  —  " 
cosrr 

T  tan'5  re  — 

cos2  re 

1 

1 

esc  re  —  •  •  •  •  . 
sin  x 

1  4"  cot  re  — 

sin2  re 

sin  (90  —  re)  =  cos  re 

sin  (-re)  = 

—  sin  re 

cos  (90  —  re)  =  sin  re 

cos  (  —  re)  = 

cos  re 

tan  (90  —  re)  =  cot  x 

tan  (  —  re)  = 

—  tan  re 

Sum  and  Difference  of  Two  Angles 

sin  (re  4-  2/) 

=  sin  re  cos  y  4-  cos  re  sin  y 

cos  (x  4-  y) 

—  cos  re  cos  y  —  sin  re  sin  y 

tan  (re  4-  y) 

tan  x  4-  tan  y 

1  —  tan  re  tan  y 

sin  (re  —  y) 

=  sin  re  cos  y  —  cos  re  sin  y 

cos  (x  —  y} 

—  cos  x  cos  y  4"  sin  re  sin  y 

tan  re  —  tan  y 

tan  (x       y) 

1  4-  tan  re  tan  y 

TRIGONOMETRIC  FORMULAS 


1-09 


Product  of  the  Functions  of  Two  Angles. 

sin  x  sin  y  =  1/2  [cos  (x  —  y)  —  cos  (x  +  y}] 
sin  x  cos  y  =  1/2  [sin  (a;  +  2/)  +  sin  (x  —  y}] 
cos  £  sin  y  =  1/2  [sin  (#  +  y)  —  sin  (a:  —  y)] 
cos  re  cos  y  —  1/2  [cos  (x  +  y)  +  cos  (a;  —  y)] 
Functions  of  Twice  an  Angle. 

sin  2x  =  2  sin  #  cos  a;         cos  2:c  =  cos2  x  —  sin2  x  =  2  cos2  a;  —  1 

2  tans; 


tan  2iX  • 


Functions  of  Half  an  Angle. 

I  —  cos  x 


1  -  tan2  x 


Functions  of  Three  Times  an  Angle. 

sin  3x  =  3  sin  x  —  4  sin3  x  cos  3rc  =  4  cos3  x  —  3  cos  x 

3  tan  x  —  tan3  x 


tan  3#  =5 


1-3  tan2  x 


TRIGONOMETRY.     Any   triangle    is    completely    denned    when  FIG.  3 

(1)  two  sides  and  the  included  angle  are  known,  (2)  one  side  and  two 
angles  are  known,  (3)  three  sides  are  known.     Let  the  sides  and  angles  of  a  triangle  be 
designated  as  in  Fig.  3. 

1.  Given  two  sides  a  and  &,  and  the  included  angle  -y.    Then 


c  =  Va2  +  62  —  2  ab  cos  7 


a    . 

-  sin  7 
c 


0  «  180  -  a  -  T 

2.  Given  the  side  a  and  the  two  angles  (3  and  7.     Then 

a  =  180  -  0  -  7 


c  —  a  • 


sm  a 
sin  7 
sin  OL 

3.  Given  the  three  sides  a,  &,  and  c.     Put 

s  =  V2  (a  +  b  +  c) 


Then 


sin 


=  r-  Vs(s  —  a)(s  — 
6c 


sin  |8  —  -  sin  a 
a 


—  c) 


•y  -  180  -  a  -  0 

Relations  between  Sides  and  Angles.    The  following  relations  between  the  sides  and 
angles  of  a  triangle  are  sometimes  useful : 

b 


a 
sin  a 


c 

sin  7 


26c 


and  similar  relations  for  the  other  two  angles. 


1-10  MATHEMATICS,   UNITS,  AND  SYMBOLS 

4.  EXPONENTIAL  AND  HYPERBOLIC  FORMULAS 

When  the  relation  between  any  variable  y  and  another  variable  x  is  such  that  x  occurs 
as  an  exponent  of  one  or  more  terms,  y  is  said  to  be  an  exponential  function  of  x.  Of 
particular  importance  in  connection  with  electric  circuits  are  the  exponential  functions 
ex  and  e~x,  where  e  is  the  base  of  the  natural  logarithms.  Since  x  is  the  natural  logarithm 
of  ex,  the  value  of  ex  can  be  obtained  from  the  table  of  common  logarithms  as  shown  at  the 
beginning  of  that  table.  In  addition  the  values  of  ex  and  e~x  are  given  in  a  separate  table. 

Hyperbolic  functions  are  an  extension  of  the  trigonometric  functions  to  those  cases 
where  the  use  of  the  latter  gives  rise  to  imaginary  or  complex  angles.  From  the  relations 


e**  -  e-** 

sin  x  —  : • 

where  j  =  v  —  1,  it  follows  that,  putting  x  —  jz: 

e*  -f  e~~*  , 

COSJZ   = (1) 

—j  sin  jz  = (2) 

Expressions  (1)  and  (2)  are  both  real  quantities  when  z  is  real,  that  is,  when  the  angle  jz 
is  imaginary.  The  first  expression  is  called  the  hyperbolic  cosine  of  z,  abbreviated  and 
pronounced  "cosh";  the  second  expression  is  called  the  hyperbolic  sine  of  z,  abbreviated 
sinh  and  pronounced  "shin."  Hence,  using  x  for  the  variable, 


sinh  x 


cosh  x  — 


2 

The  hyperbolic  tangent,  cotangent,  secant,  and  cosecant  are  defined  as  follows: 

sinh  x 


tanh  x 
coth  x  = 
sech  x  = 
csch  x  — 


cosh  a; 

cosh  x 

sinh  x 

1 


sinh  x 

The  hyperbolic  angle  a;  is  a  number  analogous  to  radians  in  circular  measure;  it  is  never 
expressed  in  degrees. 

Adding  2-Tr  to  an  angle  does  not  change  the  value  of  the  trigonometric  functions;  they 
are  therefore  said  to  have  a  period  equal  to  2?r  radians.  Hyperbolic  functions,  however, 
have  no  true  period,  but  adding  2trj  to  the  hyperbolic  angle  does  not  change  the  values 
of  the  functions;  hence  these  functions  have  an  imaginary  period,  2wj. 

For  the  value  of  the  hyperbolic  functions  see  tables  of  exponential  and  hyperbolic 
functions,  Article  13. 

Approximate  Formulas.     Note  that,  for  x  less  than  0.1, 

sinh  x  =  x  with  an  error  of  less  than  0.2  per  cent 

cosh  x  **  1  -f  —  with  an  error  of  less  than  0.09  per  cent 

£ 

For  x  greater  than  6, 

ex       1 

sinh  x  =  cosh  x  =  — •  —  -  logic"1  (0.43429#) 
2i         2i 

with  an  error  of  less  than  0.01  per  cent. 

Anti-functions.  If  a  —  sinh  x,  then  x  is  the  angle  whose  hyperbolic  sine  is  a;  this  may 
be  expressed  symbolically 

x  =  sinh"1  a 

which  is  read  "x  equals  the  angle  whose  hyperbolic  sine  is  a."    The  angle  x  is  also  called 
the  "anti-hyperbolic  sine"  or  the  "inverse  hyperbolic  sine."  of  a.    Similarly  for  the  other 


EXPONENTIAL  AND  HYPEEBOLIC  FORMULAS  1-11 

hyperbolic  functions.    The  following  relations  exist  between  the  anti-hyperbolic  functions 

and  the  natural  logarithms:  

sinh"""1  x  =  log  (x  -f  Vrc2  +  1 ) 

cosh"1  x  =  log  (x  +  Vrc2  —  1 ) 


tanh"1  x  —  -  log 

Relations  among  Functions  of  the  Same  Angle. 

cosh2  x  —  sinh2  x  —  1 

i 
1  —  tanh2  re  = 

coth2  x  -  I  = 


cosh2  x 

1 


sinh2  re 
sinh  (  —  x)  =  —  sinh  x 
cosh  ( —  x)  —  cosh  x 
tanh  (  —  x)  =  —  tanh  re 
See  also  the  definitions  given  above. 
Sum  and  Difference  of  Two  Angles.. 

sinh  (x  +  y)  —  sinh  re  cosh  y  +  cosh  re  sinh  y 
cosh  (re  -j-  2/)  =  cosh  x  cosh  y  +  sinh  re  sinh  y 

.    .      ,     .         tanh  re  -f-  tanh  y 

tanh  (re  ~r  y)  —  

1  +  tanh  x  tanh  y 

sinh  (x  —  y)  —  sinh  x  cosh  y  —  cosh  x  sinh  y 
cosh  (re  —  y}  ~  cosh  re  cosh  y  —  sinh  re  sinh  y 

,      .    .  N         tanh  x  —  tanh  y 

tanh  (x  —  y)  - — - — - — — 

1  —  tanh  x  tanh  y 

Product  of  the  Functions  of  Two  Angles. 

sinh  x  sinh  y  =  1/2  [cosh  (x  +  y)  —  cosh  (re  —  y}] 
sinh  re  cosh  y  =  1/2  [sinh  (re  +  y)  -f-  sinh  (re  —  y)] 
cosh  re  sinh  y  =  1/2  [sinh  (re  +  2/)  —  sinh  (re  —  2/)] 
cosh  re  cosh  y  —  1/2  [cosh  (#  +  2/)  -f-  cosh  (re  —  y)] 

Functions  of  Twice  an  Angle. 

sinh  2x  ~  2  sinh  re  cosh  re 

cosh  2x  =  sinh2  re  +  cosh2  re  «  2  sinh2  re  +  1  =*  2  cosh2  re  -  1 

.   rt  2  tanh  x 

tanh  2:c  =  - — -— — r-r- 
1  +  tantr  x 

Functions  of  Half  an  Angle. 


.      ^       ^  /cosh  re  —  1 
Sinn  —  — 


x       ^  /cosh  re  • 
2 


x 


2         V  Cosh  re  +  1 
Functions  of  Three  Times  an  Angle. 

sinh  3rc  =  3  sinh  re  -f-  4  sinh3  re 
cosh  3x  =  4  cosh3  #  —  3  cosh  x 

.   ^         3  tanh  x  +  tanh3  re 

tanh  3rc  =  — r^ 

1  +  3  tanh2  re 

Relations  between  Hyperbolic  and  Trigonometric  Functions. 

sinh  O'rc)  =  j  sin  re  sin  (jx)  =  y  sinh  a; 

cosh  (jx}  —  cos  re  cos  (jx)  —  cosh  re 

tanh  (jx)  ~  j  tan  re  tan  (jx)  —  j  tanh  re 

sinh^jrc  =  j  sin"1  re  sin"1^  ==  j  sinh"1  re 

tanh~xya;  =  j  tan"1  re  tan1"1^  =  j  tanh"1  x 

j  cos"1  jo;  =  log  (re 


1-12 


MATHEMATICS,   UNITS,   AND   SYMBOLS 


Hyperbolic  Functions  of  a  Complex  Angle. 

sinh  (x  +  jy]  —  sinh  x  cos  y  +  j  cosh  x  sin  3 


M 


-v 


'cosh  2x  —  cos  2y 


and     tan  0  •• 


2  tanhx 

cosh  (a;  +  jy]  —  cosh  x  cosy  +  j  sinh  re  sin  y  — 


where 


where 


where 


where 


5.  CALCULUS  FORMULAS 

The  formula  for  the  integration  by  parts  is: 

/b  j  /»& 

udv  —  [uv]    —    I   vdu 
a       Ja 

The  following  table  is  used  in  the  formulas 

#w  _  al_^ 


tan  2; 


N  = 
tanh  (x 
P  = 

A  /cosh  2x  + 

jinx-tang/ 

rj-iii- 

>                 2 
sinh 

x  cos  y  +  y  cosh  x  sin  ?/ 

cosh  x  cos  y  4-  j  sinh  x  sin  j/ 

^  /cosh  2x  — 

cos  2;y         j      i        j.        i 

I"  sin  2y  "j 

^cosh  2x  + 

f  «n>»~l 

cos  2y 
4.  cos  o;~l                            1 

ww  ~.            o-nrl         R      -  -         i- 

Lsinh2xJ 

2  tann      1    j_ 

jo         ana     z>2  —  rt  t 
H-  A2  J                          2 

'an      [l-  ^L2 

^ 


/(a?)  + 


where  C  is  an  arbitrary  constant. 


/'(x) 

/(*) 

/'(*> 

/W 

1 

1 

1 

X 

m  +  1  ^  . 

cos2  ax 

a 

I 

^ 

1 

1 

ax 

a       e  ^ 

sin2  ax 

a 

, 

1 

1                             y  X 

-,ax 

mn  -1  V        7i 

a 

Vaa  +  bx2 

bx 

1        6a. 

1 

1                   _,   ^/—  1 

b  log  a  "" 

xVx2  +  a 

1    . 

x 

±-\/^2  _L.  ,>.2 

a 

Va*  ±  x- 

1 

x 

V~2  2 

a 

Vx2  -  a2 

_-_'L1     ~_ 

!    . 

dx          dx 

V 

cosn  ax 

a 

uz 

U 

sinh  ax 

-  cosh  ax 

log  x 

x  log  x  —  x 

a 

tan  ax 

log  (cos  ax) 

sin2x 

—  1/a(cos  x  sin  x  —  x) 

a 

taxxh  ax 

-  log  (cosh  ax) 

COS2X 

1/2  (sin  x  cosx-}~  x) 

a 

DIFFERENTIAL  EQUATIONS  1-13 

MAXIMA  AND  MINIMA.     Let  y  be  any  function  of  a  variable  re;  then  y  will  be  a 
maximum  or  minimum  for  any  value  of  x  which  satisfies 


(1) 

u>u/ 

provided  -r-^  is  not  zero.    If  the  second  derivative  ~~  is  positive  for  this  value  of  x,  then 

the  corresponding  value  of  y  is  a  minimum;  if  this  second  derivative  is  negative,  the  cor- 
responding value  of  y  is  a  maximum. 

In  case  —^  is  also  zero  for  the  value  of  x  which  satisfies  (1),  the  corresponding  value  of 

y  is  not  a  maximum  or  minimum  unless  -~  is  also  zero  and  —  is  not  zero.    When  —  —  0, 

dx*  dx*  dx3 

.   .  d*y  .          .  .  cfiy  d*y 

2/  is  a  minimum  if  — -.  is  positive  and  a  maximum  if  — -  is  negative.    In  case  — ;  is  also  zero, 

ax  ax  dx 

similar  relations  must  hold  for  the  fifth  and  sixth  derivatives,  etc. 

6.  DIFFERENTIAL  EQUATIONS 

Differential  equations  of  the  following  forms  are  met  with  in  the  theory  of  alternating 
and  transient  currents. 

The  following  notation  is  used:  e  =  2.7183-  •  •  =  base  of  natural  system  of  logarithms; 
x,  y,  z  are  variables.  A,  <j>,  7,  and  $  are  constants  of  integration  or  arbitrary  constants. 
Other  letters  represent  known  constants. 

Ay  f\\ 

-—  =  ay  (1) 

dx  ' 

Solution:  y  =  Aeax 

—  -j-  ay  =  0  (2) 

Solution:  y  =  Ae~ax 

^  +  ay  «  b  (3) 

dx 

Solution:  y  =  -  [1  —  Ae~ax] 

a 

d2y 
Solution:  y  =  A  sin  (arc  +  <£) 

-7-5  =  a2?/  (5) 

Solution:  y  =  A  sinh  (ax  +  4>) 

Solution : 

Case     I.  a2  positive:    y  —  Ae~ux  sin  (ax  +  <£) 

Case    II.  a2  negative:  y  =  Ae~^x  sinh  (ax  +  <£) 

Case  III.  a2  =  0:  y  =  A(x  +  <^e-^x 

--T  •{•  2u- — h  (M?  +  az)y  —  B  sin  (cox  +  0)  (7) 

aar  dx  , 

The  complete  solution  of  this  equation  consists  of  the  solution  of  (6)  plus  the  term 

I  sin  (ux  +  $  —  5)  (a) 

where  5  =  tan"1         ""T -2 

O     T"  W     —    W 

For  each  additional  sine  term  added  to  the  right-hand  member  of  the  equation,  there 
will  be  a  corresponding  term  of  the  same  form  as  (a)  in  the  solution. 


(B  sin  5\    . 

I  — J  sm 

\    2ua>    / 


1-14  MATHEMATICS,  TJNITS,  AND  SYMBOLS 

B  •*»<»*  +  «>  (8) 


Solution: 

y  =  Aiemi*  +  A->em*x  4  •  -  •  ^n^771**  +  •£#  sin 


where  wii,  W2,  etc  ,  are  the  n  roots  of  the  equation 

mn  4-  On-ittt71"1  4  ----  aim  +  OQ  =  0 

and  JT  and  5  are  found  by  substituting  the  JTS  sin  (tax  +  0  4  5)  by  itself  in  the  given 
differential  equation  and  equating  the  coefficients  of  sin  (ojrc  +  6}  and  cos  (W  -f-  0) 
respectively  on  the  two  sides  of  the  resulting  equation.  When  the  second  member  of 

the  differential  equation  is  a  constant,  B,  the  sine  term  in  the  solution  becomes  simply  —  . 
Note  that  all  the  preceding  equations  are  merely  special  cases  of  the  general  equation  (8) 


The  complete  solution  of  this  equation  contains  an  infinite  number  of  terms  of  the  form 
y  =  e-(*-»)*  [Aie™  sin  (wx  +  nz  +  <£i)  +  A^e~mz  sin  (ux  —  nz  +  <#>2)]  (a) 

where  Ai,  <£i,  A+,  <fa,  and  two  of  the  four  constants  «,  s,  m,  and  n  are  integration  constants 
(fixed  by  the  terminal  conditions).    The  values  of  m  and  n  in  terms  of  o>  and  s  are 

/-r        ^  +  e 
in  —  cV  ab  cos  —  -  — 
2 

.  /-T   .     17  4-  e 
n  =  c  V  ao  sin  —  -  — 

where  a  =  V(s  +  g)2  +  co2,  e  =  tan~J  (  —  ^—  ) 

\«  +  «/ 


6  =  V(s  -  e)2  +  w2,  77  =  tan 

The  values  of  co  and  s  in  terms  of  m  and  n  are 

VJ'G        a  +  0 


__ 

where  F  «  V(n  +  c^)2  +  m2,  a  =  tan"1 

G  =  V(n  -  cg)2  4  m2,  0  =  tan~i 


n 


The  solution  of  eq.  (9)  may  also  be  written  as  a  series  of  terms  of  the  form 

y  =  Me~(u~*)x  sin  (ux  4  <£  4  /u)  (6) 

•W  =  -p=  Vcosh  2,(mz  4  7)  4  cos  2 (712  4  0) 

tan  /t  =  tanh  (mz  4-  7)  tan  (nz  4  0) 

where  A,  <$>,  7,  and  0  are  integration  constants,  and  the  relations  between  the  other  con- 
stants 6>,  s,  m,  and  n  are  the  same  as  above.  * 

In  the  special  case  when  q  =  0,  the  solution  of  eq.  (9)  is 


y  =  fl-tw  jy^ua;  4.  n 
where /i  and/2  are  any  two  arbitrary  functions  and  u  and  n  are  connected  by  the  relation 


ERRORS   OF  OBSERVATION  1-15 


=  0  (10) 

This  is  known  as  Bessel's  equation  of  order  n.  Jn(x),  Bessel's  function  of  the  first  kind  of 
order  n,  is  a  particular  solution  of  this  equation.  It  may  be  computed  from  the  infinite 
series : 

xn          f  x2  ~4 

J    (x\    __ I    1    .._... L, 


2nr(w  -f  1)  L          22(n  +  1)        242!(n  +  l)(n  +  2) 

2*3Kn +  !)(*'+ 2)(»  + 3)  +• 

where  T(n  +  1)  is  the  gamma  function  which  reduces  to  unity  for  n  =  0  and  to  nl  for  n 
equal  to  any  positive  integer.  In  general,  the  function  Jn(x)  is  an  oscillatory  function  of 
x  having  the  value  zero  for  x  =  0,  except  for  the  case  where  n  —  0.  For  values  of  n  larger 
than  1,  the  slope  of  Jn(x)  is  zero  for  x  —  0  and  the  first  maximum  and  the  first  zero  occurs 
at  successively  higher  values  of  x  as  n  takes  on  larger  values.  For  small  values  of  n,  the 
values  of  x  for  which  Jn(x}  is  a  maximum  or  zero  can  be  gotten  from  tables  of  Bessel  func- 
tions. For  large  values  of  n,  the  first  maximum,  that  is,  the  smallest  x  for  which  Jn'(x} 
—  0,  is  given  by 

n  +  0.809  v/n  (b) 

with  an  error  not  larger  than  l/^/n,  and  the  first  zero,  that  is,  the  smallest  x  for  which 
Jn(x)  =  0,  is  given  by 

n  -}-  1.856  -v/n  (e) 

again  with  an  error  of  the  order  of  l/\/n. 
For  integral  values  of  n  greater  than  zero, 

2n 
x 

which  permits  one  to  compute  Bessel  functions  for  successively  higher  order  from  tables 
of  JQ(X)  and  Ji(x). 
When  n  is  an  integer, 


BIBLIOGRAPHY 

Watson,  G.  N.,  A  Treatise  on  the  Theory  of  Bessel  Functions,  2nd  Ed.,  New  York,  The  Macmillan  Com- 
pany (1944). 
Article  14  of  this  section. 

7.  ERRORS  OF  OBSERVATION 

When  a  quantity  is  measured  with  all  possible  accuracy  many  times  in  succession,  the 
numbers  expressing  the  results  are  found  to  differ  by  amounts  which,  although  generally 
small,  are  occasionally  considerable  in  comparison  with  the  quantity  measured.  Though 
these  differences  may  be  decreased  by  improved  methods,  better  instruments,  or  greater 
skill,  they  can  never  be  entirely  removed.  They  are  known  as  the  errors  of  observation. 
The  following  formulas,  which  are  derived  from  the  theory  of  least  squares,  apply  to  such 
errors  and  not  to  errors  which  can  be  eliminated  by  correcting  mistakes  of  the  observer  or 
defects  of  instruments  or  methods  of  observation.  That  is,  they  apply  only  to  errors 
which  may  be  either  positive  or  negative,  the  chance  of  a  positive  error  occurring  being 
exactly  the  same  as  the  chance  of  a  negative  error  occurring. 

WEIGHTED  OBSERVATIONS.  Sometimes,  in  spite  of  the  care  with  which  obser- 
vations are  taken,  there  are  reasons  for  believing  that  some  observations  are  better  than 
others.  In  this  case  the  observations  are  given  different  "weights"  or  numbers  express- 
ing their  relative  practical  worth.  A  weighted  observation  is  an  observation  multiplied 
by  its  weight. 

PROBABLE  VALUE  OF  SEVERAL  OBSERVATIONS.  The  most  probable  value  of 
a  quantity  which  is  observed  directly  several  times  with  equal  care  is  the  arithmetical 
mean  of  the  measurements. 

The  most  probable  value  of  a  quantity  which  is  observed  directly  several  times,  but 
the  observations  of  which  have  different  weights,  is  equal  to  the  sum  of  the  weighted 
observations  divided  by  the  sum  of  the  weights. 


1-16  MATHEMATICS,  UNITS,  AND  SYMBOLS 

PROBABLE  ERROR  OF  ANY  ONE  OF  SEVERAL  OBSERVATIONS.  The  probable 
error  or  dispersion  of  a  number  of  direct  observations  made  with  equal  care  is  given  by 
the  following  formula: 


0.6745 


*  n  —  1 
where  n  =  number  of  observations. 

r  -  probable  error  of  a  single  observation. 

0  =  residual  found  by  subtracting  the  arithmetical  mean  from  each  measurement. 

The  probable  error  of  each  of  a  number  of  direct  observations,  where  the  observations 
have  different  weight,  is  found  by  the  following  formula,  in  which  p  represents  the  per 
unit  weight  of  an  observation. 


0.6745 


«n  -  1 
PROBABLE  ERROR  OF  THE  ARITHMETICAL  MEAN.    If 

r  =  probable  error  of  a  single  observation, 

n  =  number  of  observations, 

TQ  =  probable  error  of  the  arithmetical  mean, 

TQ  =  — —  for  observations  of  equal  weight 

Vn 
or 

Tt 

:  for  unequal  weight 


It  should  be  noted  that  the  probable  error  of  the  mean  decreases  inversely  as  the  square 
root  of  the  number  of  observations. 

PROBABLE  ERROR  IN  A  RESULT  CALCULATED  FROM  THE  MEANS  OF  SEV- 
ERAL OBSERVED  QUANTITIES.  Let  Z  «  a  sum  or  difference  of  several  independent 
quantities. 

Let  ri,  72,  r3,  etc.,  be  the  probable  errors  in  these  quantities.  Then  the  probable  error  of 
Z  is  equal  to 

Wi2  +  r22  +  r32  +  etc. 

Let  Z  —  Az,  where  z  is  an,  observed  quantity,  and  A,  a  known  number.  Let  r  be  the 
probable  error  in  z.  Then  the  probable  error  in  Z  is  Ar. 

Let  Z  be  the  product  of  two  independently  observed  quantities  z\  and  22  whose  probable 
errors  are  r\  and  r%  respectively.  Then  the  error  in  Z  is  equal  to 


4- 

Let  Z  be  any  function  of  the  independently  observed  quantities  z\,  zz,  23,  etc.,  whose 
probable  errors  are  ri,  r^  TZ,  etc.    Then  the  probable  error  in  Z  is  equal  to 


8.  APPROXIMATIONS 

If  a  is  small 

(1  d=  a) m  =  1  ±  ma 
If  mis  nearly  equal  to  n 

m  -f-  n 


If  #,  expressed  in  radians,  is  small  compared  to  a  radian 

sin  0  =  tan  8  =  6  radians 


MENSURATION  1-17 

9.  SERIES 


Taylor's  series  is  written 


f(x  +  K)  =  /(a»  -f 


where  the  prime  on  the  function  means  the  derivative  with  respect  to  the  argument. 
The  following  series  are  frequently  useful. 

#2          £.3 

**  =  !+*  +  -  +  -+... 


a3    ,   z6       x     . 
*-  -  +  ---+.- 

0^         X*         X6 

cos*-  1-55  +  ---+..- 

cos  (a;  sin  0)  =  /0(a:)  +  2  {/2(»  cos  20  +  /4(aj)  cos  4^  H  ---- 
where  Jn(x)  is  Bessel's  function  of  order  n, 

sin  (x  sin  0)  =  2  {«/i(z)  sin  0  +  ^(a;)  sin  30  H  ---- 
cos  (x  cos  0)  =  «70(a;)  -  2/2(35)  cos  29  +  2/4(0:)  cos  4:6  -\  ---- 
sin  (x  cos  0)  =  2/i(z)  cos  0  —  2/3(2)  cos  30  +  2/6(aO  cos  50  H  ---- 
sin  (A  +  x  sin  0)  =  JQ(x)  sin  A  +  /i  (re)  [sin  (-4.  +  0)  -  sin  (A  -  0)] 
+  /a(aO[sin  (A  +  20)  +  sin  (A  -  20)] 
4-  /s(a:)[sin  (A  +  30)  -  sin  (A  -  30)] 
+  /4(aO[sin  (A  +  40)  +  sin  (A  -  40)]  H  ---- 

10.  MENSURATION 

The  term  mensuration  is  used  in  this  article  to  include  the  relations  between  the  areas 
and  volumes  of  geometric  figures  and  their  linear  dimensions. 
Triangle. 

Area  =  1/2  (Base)  X  (Perpendicular  height) 


=  Vs(s  —  a)(s  —  ty(s  —  c) 

where  a,  6,  and  c  are  the  lengths  of  the  three  sides  respectively,  and  s  —  1/2  (a  +  6  +  c). 
Trapezoid. 

Area  -  (  ^-^  )  d 


where  a  and  6  are  the  lengths  of  the  parallel  sides  respectively,  and  d  their  distance  apart. 

Parallelogram. 

Area  =  (Base)  X  (Perpendicular  height) 
Parabola. 

Area  =  2/3  (Area  of  circumscribing  rectangle) 
Cycloid. 

Area  =  3  UTTX  (Altitude)2 

the  altitude  being  the  diameter  of  the  rolling  circle. 


1-18  MATHEMATICS,  UNITS,  AND  SYMBOLS 

Circle. 


Circumference  =  2?rr  =  ird 

1 
where  r  is  the  radius  and  d  the  diameter. 


Area  =  xr2  =  —  < 
4 


Area  of  segment  =  —  (6  —  sin  0) 

A 


where  6  is  the  angle  in  radians  (see  Angles)  subtended  by  the  arc  of  the  segment.    If  n  is 
the  height  of  the  segment,  measured  along  the  radius  perpendicular  to  the  chord, 


where 
Ellipse. 


Area  of  segment  =  irfi  M  —  A(r  —  ri) 


Area  =  irab 


where  a  and  6  are  the  principal  semi-axes. 
Prism  with  Parallel  Sides  and  Parallel  Ends. 

Volume  =  (Area  of  end)  X  (Perpendicular  distance  between  ends) 
Right  Circular  Cylinder. 

Volume  =  ~  d2  1 
4 

where  d  is  the  diameter  and  I  the  length. 

Total  surface  of  right  cylinder  =  ird(l  +  l/%d) 
Right  Circular  Cone. 

Volume  =  1/3  (Area  of  base)  X  (Height) 

=  !/3  (Volume  of  circumscribing  cylinder) 
where  r  is  the  radius  of  base  and  h  the  height  of  the  cone. 

Area  of  curved  surface  of  a  right  circular  cone  =  irr  Vhz  +  r2 
Right  Pyramid. 

Volume  =  1/3  (Area  of  base)  X  (Height). 

Volume  of  frustum  of  pyramid  =  1/3  (Height)  (A  +  VaA  ) 

where  A  and  a  are  the  areas  of  the  ends  respectively. 
Sphere. 

T  —  radius 

Area  of  surface  =  4irr*  —  2/3  (total  area  of  circumscribing  cylinder) 

Area  of  the  surface  of  a  zone  of  a  sphere  «  area  of  zone  of  the  same  height  as  this  zone 
projected  on  to  a  cylinder. 

Volume  =  4/3  xT-3  =  2/3  (volume  of  circumscribing  cylinder) 

Volume  of  a  frustum  of  a  sphere  =  ^(k  ±h)  —  -  (kz  ±  A3),  where  k  is  the  distance 

3 

of  its  outer  face  from  center  and  Ji  the  distance  of  its  inner  face  from  the  center,  the  nega- 
tive signs  in  the  brackets  to  be  used  if  both  faces  are  on  the  same  side  of  the  center  and 
the  positive  signs  if  on  opposite  sides  of  the  center. 
Ellipsoid. 

Volume  = 


where  a,  &,  and  c  are  the  three  principal  semi-axes,  respectively. 

Paraboloid.    Volume  of  a  paraboloid  of  revolution  equals  one-half  that  of  the  circum- 
scribing cylinder* 


COMMON  AND  NATURAL  LOGARITHMS  OF  NUMBERS      1-19 


MATHEMATICAL  TABLES  AND  CHARTS 


11.    COMMON  AND  NATURAL  LOGARITHMS  OF  NUMBERS 

The  common,  logarithm  of  a  number  is  the  index  of  the  power  to  which  the  base  10 
must  be  raised  in  order  to  equal  the  number. 

The  common  logarithm  of  every  positive  number  not  an  integral  power  of  10  consists 
of  an  integral  and  a  decimal  part.  The  integral  part  or  whole  number  is  called  the  charac- 
teristic and  may  be  either  positive  or  negative.  The  decimal  or  fractional  part  is  a  positive 
number  called  the  mantissa  and  is  the  same  for  all  numbers  which  have  the  same  sequential 
digits. 

The  characteristic  of  the  logarithm  of  any  positive  number  greater  than  one  is  positive 
and  is  one  less  than  the  number  of  digits  before  the  decimal  point. 

The  characteristic  of  the  logarithm  of  any  positive  number  less  than  one  is  negative 
and  is  one  more  than  the  number  of  ciphers  immediately  after  the  decimal  point. 

A  negative  number  or  number  less  than  zero  has  no  real  logarithm. 

EXAMPLES:    Logio  25400.  =  4,404834    Logio  0.0254  =  2.404834  or  8.404834  —  10 

The  two  systems  of  logarithms  in  general  use  are  the  common  or  Briggsian  logarithms, 
introduced  in  1615  by  Henry  Briggs,  a  contemporary  of  John  Napier,  the  inventor  of 
logarithms,  and  the  natural  or  less  appropriately  termed  Napierian  or  hyperbolic  loga- 
rithms, which  developed  somewhat  accidentally  from  Napier's  original  work.  The  latter 
have  a  base  denoted  by  e,  an  irrational  number,  which  is: 


2.7182818 


To  obtain  the  natural  logarithm,  the  common  logarithm  given  below  is  multiplied 
by  loge  10  which  is  2.302585,  or  log,  N  =  2.302585  logio  N. 


N 

0 

1 

2 

3 

4 

5 

6 

7 

8 

9 

0 

000000 

301030 

477121 

602060 

698970 

778151 

845098 

903090 

954243 

1 

2 
3 

000000 
301030 
477121 

041393 
322219 
491362 

079181 
342423 
505150 

113943 
361728 
518514 

146128 
380211 
531479 

176091 
397940 
544068 

204120 
414973 
556303 

230449 
431364 
568202 

255273 
447158 
579784 

278754 
462398 
591065 

4 
5 
6 

602060 
698970 
778151 

612784 
707570 
785330 

623249 
716003 
792392 

633468 
724276 
799341 

643453 
732394 
806180 

653213 
740363 
812913 

662758 
748188 
819544 

672098 
755875 
826075 

681241 
763428 
832509 

690196 
770852 
838849 

7 

8 
9 

845098 
903090 
954243 

851258 
908485 
959041 

857332 
913814 
963788 

863323 
919078 
968483 

869232 
924279 
973128 

875061 
929419 
977724 

880814 
934498 
982271 

886491 
939519 
9867.72 

892095 
944483 
991226 

897627 
949390 
995635 

10 

000000 

004321 

008600 

012837 

017033 

021189 

025306 

029384 

033424 

037426 

1 
2 
3 

041393 
079181 
113943 

045323 
082785 
117271 

049218 
086360 
120574 

053078 
089905 
123852 

056905 
093422 
127105 

060698 
096910 
130334 

064458 
100371 
133539 

068186 
103804 
136721 

071882 
107210 
139879 

075547 
110590 
143015 

4 
5 
6 

146128 
176091 
204120 

149219 
178977 
206826 

152288 
181844 
209515 

155336 
184691 
212188 

158362 
187521 
214844 

161368 
190332 
217484 

164353 
J  93  125 
220108 

167317 
195900 
222716 

170262 
198657 
225309 

173186 
201397 
227887 

7 
8 
9 

230449 
255273 
278754 

232996 
257679 
281033 

235528 
260071 
283301 

238046 
262451 
285557 

240549 
264818 
287802 

243038 
267172 
290035 

245513 
269513 
292256 

247973 
271842 
294466 

250420 
274158 
296665 

252853 
276462 
298853 

20 

301030 

303196 

305351 

307496 

309630 

311754 

313867 

315970 

318063 

320146 

1 
2 
3 

322219 
342423 
361728 

324282 
344392 
363612 

326336 
346353 
365488 

328380 
348305 
367356 

330414 
350248 
369216 

332438 
352183 
371068 

334454 
354108 
372912 

336460 
356026 
374748 

338456 
357935 
376577 

340444 
359835 
378398 

4 
5 
6 

380211 
397940 
414973 

382017 
399674 
416641 

383815 
401401 
418301 

385606 
403121 
419956 

387390 
404834 
421604 

389166 
406540 
423246 

390935 
408240 
424882 

392697 
409933 
426511 

394452 
41  1620 
428135 

396199 
413300 
429752 

7 
8 
9 

431364 
447158 
462398 

432969 
448706 
463893 

434569 
450249 
465383 

436163 
451786 
46686S 

437751 
453318 
468347 

439333 
454845 
469822 

440909 
456366 
471292 

442480 
457882 
472756 

444045 
459392 
474216 

445604 
460898 
475671 

30 

477121 

478566 

480007 

481443 

482874 

484300 

485721 

487133 

438551 

489958 

1 
2 
3 

491362 
505150 
518514 

492760 
506505 
519828 

494155 
507856 
521138 

495544 
509203 
522444 

496930 
510545 
523746 

498311 
511883 
525045 

499687 
513218 
526339 

501059 
514548 
527630 

502427 
515874 
528917 

503791 
517196 
530200 

4 
5 

531479 
544068 

532754 
545307 

534026 
546543 

535294 
547775 

536558 
549003 

537819 
550228 

539076 
551450 

540329 
552668 

541579 
553883 

542825 
555094 

1-20 


MATHEMATICS,  UNITS,  AND  SYMBOLS 


N 

0 

1 

2 

3 

4 

5 

6 

7 

8 

9 

P-- 
j 

544068 

545307 

546543 

547775 

549003 

550228 

551450 

552668 

553883 

555094 

6 

556303 

557507 

558709 

559907 

561101 

562293 

563481 

564666 

565848 

567026 

7 

568202 

569374 

570543 

571709 

572872 

574031 

575188 

576341 

577492 

578639 

8 

579784 

580925 

582063 

583199 

584331 

585461 

586587 

587711 

588832 

589950 

c 

591065 

592177 

593286 

594393 

595496 

596597 

597695 

598791 

599883 

600973 

40 

602060 

603144 

604226 

605305 

606381 

607455 

608526 

609594 

610660 

611723 

1 

612784 

613842 

614897 

615950 

617000 

618048 

619093 

620136 

621176 

622214 

623249 

624232 

625312 

626340 

627366 

628389 

629410 

630428 

631444 

632457 

a 

633468 

634477 

635484 

636488 

637490 

638489 

639486 

640481 

641474 

642465 

^ 

643453 

644439 

645422 

646404 

647383 

648360 

649335 

650308 

651278 

652246 

e 

653213 

654177 

655138 

656098 

657056 

658011 

658965 

659916 

660865 

661713 

6 

662758 

663701 

664642 

665581 

666518 

667453 

668386 

669317 

670246 

671173 

7 

672098 

673021 

673942 

674861 

675778 

676694 

677607 

678518 

679428 

680336 

8 

681241 

682145 

683047 

683947 

684845 

685742 

686636 

687529 

688420 

689309 

9 

690196 

691081 

691965 

692847 

693727 

694605 

695482 

696356 

697229 

698100 

50 

698970 

699838 

700704 

701568 

702431 

703291 

704151 

705008 

705864 

706718 

1 

707570 

708421 

709270 

710117 

710963 

711807 

712650 

713491 

714330 

715167 

2 

716003 

716838 

717671 

718502 

719331 

720159 

720986 

721811 

722634 

723456 

724276 

725095 

725912 

726727 

727541 

728354 

729165 

729974 

730782 

731589 

4 

732394 

733197 

733999 

734800 

735599 

736397 

737193 

737987 

738781 

739572 

5 

740363 

741152 

741939 

742725 

743510 

744293 

745075 

745855 

74663' 

747412 

6 

748188 

748963 

749736 

750508 

751279 

752048 

752816 

753583 

754348 

755112 

7 

755875 

756636 

757396 

758155 

758912 

759668 

760422 

761176 

761928 

762679 

8 

763428 

764176 

764923 

765669 

766413 

767156 

767898 

768638 

769377 

770115 

9 

770852 

771587 

772322 

773055 

773786 

774517 

775246 

775974 

776701 

777427 

€0 

778161 

778874 

779596 

780317 

781037 

781755 

782473 

783189 

783904 

784617 

1 

785330 

786041 

786751 

787460 

788168 

788875 

789581 

790285 

790988 

791691 

2 

792392 

793092 

793790 

794488 

795185 

795880 

796574 

797268 

797960 

798651 

3 

799341 

800029 

800717 

801404 

802089 

802774 

803457 

804139 

804821 

805501 

4 

806180 

806858 

807535 

80821  1 

808886 

809560 

810233 

810904 

811575 

812245 

5 

812913 

813581 

814248 

814913 

815578 

816241 

816904 

817565 

818226 

818885 

6 

819544 

820201 

820858 

821514 

822168 

822822 

823474 

824126 

824776 

825426 

7 

826075 

826723 

827369 

828015 

828660 

829304 

829947 

830589 

831230 

831870 

8 

832509 

833147 

833784 

834421 

835056 

835691 

836324 

836957 

837588 

838219 

9 

838849 

839478 

840106 

840733 

841359 

841985 

842609 

843233 

843855 

844477 

70 

845098 

845718 

846337 

846955 

847573 

848189 

848805 

849419 

850033 

850646 

I 

851258 

851870 

852480 

853090 

853698 

854306 

854913 

855519 

856124 

856729 

2 

857332 

857935 

858537 

859138 

859739 

860338 

860937 

861534 

862131 

862728 

3 

863323 

863917 

864511 

865104 

865696 

866287 

866878 

867467 

868056 

868644 

4 

869232 

869818 

870404 

870989 

871573 

872156 

872739 

873321 

873902 

874482 

5 

875061 

875640 

876218 

876795 

877371 

877947 

878522 

879096 

879669 

880242 

6 

880814 

881385 

881955 

882525 

883093 

883661 

884229 

884795 

885361 

885926 

7 

886491 

887054 

887617 

888179 

888741 

889302 

889862 

890421 

890980 

891537 

8 
9 

892095 
897627 

892651 
898176 

893207 
898725 

893762 
899273 

894316 
899821 

894870 
900367 

895423 
900913 

895975 
901458 

896526 
902003 

897077 
902547 

30 

903090 

903633 

904174 

904716 

905256 

905796 

906335 

908874 

907411 

907949 

1 
2 
3 

908485 
913814 
919078 

909021 
914343 
919601 

909556 

914872 
920123 

910091 
915400 
920645 

910624 
915927 
921166 

911158 
916454 
921686 

911690 
916980 
922206 

912222 
917506 
922725 

912753 
918030 
923244 

913284 
918555 
923762 

•4 
•6 

924279 
929419 
934498 

924796 
929930 
935003 

925312 
930440 
935507 

925828 
930949 
936011 

926342 
931458 
936514 

926857 
931966 
937016 

927370 
932474 
937518 

927883 
932981 
938019 

928396 
933487 
938520 

928908 
933993 
939020 

8 
9 

939519 
944483 
949390 

940018 
944976 
949878 

940516 
945469 
950365 

941014 
945961 
950851 

941511 
946452 
951338 

942008 
946943 
951823 

942504 
947434 
952308 

943000 
947924 
952792 

943495 
948413 
953276 

943989 
948902 
953760 

to 

954243 

954725 

955207 

955688 

956168 

956649 

957128 

957607 

958086 

958564 

1 
3 

959041 
963788 
968483 

959518 
964260 
968950 

959995 
964731 
969416 

960471 
965202 
969882 

960946 
965672 
970347 

961421 
966142 
970812 

961895 
966611 
971276 

962369 
967080 
971740 

962843 
967548 
972203 

963316 
968016 
972666 

4 
6 

973128 
977724 
982271 

973590 
978181 
982723 

974051 
978637 
983175 

974512 
979093 
983626 

974972 
979548 
984077 

975432 
980003 
984527 

975891 
980458 
984977 

976350 
980912 
985426 

976808 
981366 
985875 

977266 
981819 
986324 

7  , 
8 
9 

986772 
991  226 
995635 

987219 
991669 
996074 

987666 
992UI 
996512 

988113 
992554 
996949 

988559 
992995 
997386 

989005 
993436 
997823 

989450 
993877 
998259 

989895 
994317 
998695 

990339 
994757 
999131 

990783 
995196 
999565 

100 

000000 

000434 

000368 

001301 

001734 

002166 

002598 

003029 

003461 

003891 

TRIGONOMETRIC  TABLES 


1-21 


12.    TRIGONOMETRIC  TABLES 

The  following  tables  give  the  values  of  sin  x,  cos  x,  and  tan  x  for  values  of  x  from  0  to 
90°  in  intervals  of  0.1  degree.  By  making  use  of  the  periodic  character  of  these  functions, 
the  values  can  be  determined  from  these  tables  for  all  values  of  x  to  an  accuracy  of  0.1 
degree.  (See  Trigonometric  Formulas.) 

1 5^n 

If  the  angle  is  given  in  radians  multiply  the  number  of  radians  by (57.295)   to 

7T 

obtain  the  number  of  degrees. 


Trigonometric  Functions 


0.0°-15.9° 


Angle 
in 
Degrees 

Name 
of 
Function 

Value  of  Function  for  Each  Tenth  of  a  Degree 

0.0 

0.1 

0.2 

0.3 

0.4 

0.5 

0.6 

0.7 

0.8 

0.9 

sin 

0.0000 

0.0017 

0.0035 

0.0052 

0.0070 

0.0087 

0.0105 

0.0122 

0.0140 

0.0157 

0 

C03 

1.0000 

1.0000 

1.0000 

1.0000 

1.0000 

1.0000 

0.9999 

0.9999 

0.9999 

0.9999 

tan 

0.0000 

0.0017 

0.0035 

0.0052 

0.0070 

0.0087 

0.0105 

0.0122 

0.0140 

0.0157 

sin 

0.0175 

0.0192 

0.0209 

0.0227 

0.0244 

0.0262 

~0.0279 

0.0297 

0.0314 

0.0332 

1 

cos 

0.9998 

0.9998 

0.9998 

0.9997 

0.9997 

0.9997 

0.9996 

0.9996 

0.9995 

0.9995 

tan 

0.0175 

0.0192 

0.0209 

0.0227 

0.0244 

0.0262 

0.0279 

0.0297 

0.0314 

0.0332 

sin 

0.0349 

0.0366 

0.0384 

0.0401 

0.0419 

0.0436 

0.0454 

0.0471 

0.0488 

0.0506 

2 

cos 

0.9994 

0.9993 

0.9993 

0.9992 

0.9991 

0.9990 

0.9990 

0.9989 

0.9988 

0.9987 

tan 

0.0349 

0.0367 

0.0384 

0.0402 

0.0419 

0.0437 

0.0454 

0.0472 

0.0489 

0.0507 

sin 

0.0523 

0.0541 

0.0558 

0.0576 

0.0593 

0.0610 

0.0628 

0.0645 

0.0663 

0.0680 

3 

cos 

0.9986 

0.9985 

0.9984 

0.9983 

0.9982 

0.9981 

0.9980 

0.9979 

0.9978 

0.9977 

tan 

0.0524 

0.0542 

0.0559 

0.0577 

0.0594 

0.0612 

0.0629 

0.0647 

0.0664 

0.0682 

sin 

0.0698 

0.0715 

0.0732 

0.0750 

0.0767 

0.0785 

0.0802 

0.0819 

0.0837 

0.0854 

4 

cos 

0.9976 

0.9974 

0.9973 

0.9972 

0.9971 

0.9969 

0.9968 

0.9966 

0.9965 

0.9963 

tan 

0.0699 

0.0717 

0.0734 

0.0752 

0.0769 

0.0787 

0.0805 

0.0822 

0.0840 

0.0857 

sin 

0.0872 

0.0889 

0.0906 

0.0924 

0.0941 

0.0958 

0.0976 

0.0993 

0.1011 

0.1028 

5 

cos 

0.9962 

0.9960 

0.9959 

0.9957 

0.9956 

0.9954 

0.9952 

0.9951 

0.9949 

0.9947 

tan 

0.0875 

0.0892 

0.0910 

0.0928 

0.0945 

0.0963 

0.0981 

0.0998 

0.1016 

0.1033 

sin 

0.1045 

0.1063 

0.1080 

0.1097 

0.1115 

0.1132 

0.1149 

0.1167 

0.1184 

0.1201 

6 

cos 

0.9945 

0.9943 

0.9942 

0.9940 

0.9938 

0.9936 

0.9934 

0.9932 

0.9930 

0.9928 

tan 

O.I05I 

0.1069 

0.1086 

0.1104 

0.1122 

0.1139 

0.1157 

0.1175 

0.1192 

0.1210 

sin 

0.1219 

0.1236 

0.1253 

0.1271 

0.1288 

0.1305 

0.1323 

0.1340 

0.1357 

0.  1374 

7 

cos 

0.9925 

0.9923 

0.9921 

0.9919 

0.9917 

0.9914 

0.9912 

0.9910 

0.9907 

0.9905 

tan 

0.1228 

0.1246 

0.1263 

0.1281 

9.1299 

0.1317 

0.1334 

0.1352 

0.1370 

0.1388 

sin 

0.1392 

0.1409 

0.1426 

0.1444 

0.1461 

0.1478 

0.1495 

0.1513 

0.1530 

0.1547 

8 

cos 

0.9903 

0.9900 

0.9898 

0.9895 

0.9893 

0.9890 

0.9888 

0.9885 

0.9882 

0.9880 

tan 

0.1405 

0.1423 

0.1441 

0.1459 

0.1477 

0.1495 

0.1512 

0.1530 

0.1548 

0.1566 

sin 

0.1564 

0.1582 

0.1599 

0.1616 

0.1633 

0.1650 

0.1663 

0.1685 

0.1702 

0.1719 

9 

cos 

0.9877 

0.9874 

0.9871 

0.9869 

0.9866 

0.9863 

0.9860 

0.9857 

0.9854 

0.9851 

tan 

0,1584 

0.1602 

0.1620 

0.1638 

0.1655 

0.1673 

0.1691 

0.1709 

0.1727 

0.1745 

sin 

0.1736 

0.1754 

O.I77I 

0.1788 

0.1805 

0.1822 

0.1840 

0.1857 

0.1874 

0.1891 

10 

cos 

0.9848 

0.9845 

0.9842 

0.9839 

0.9836 

0.9833 

0.9829 

0.9826 

0.9823 

0.9820 

tan 

0.1763 

0.1781 

0.1799 

0.1817 

0.1835 

0.1853 

0.1871 

0.1890 

0.1908 

0.1926 

sin 

0.1908 

0.1925 

0.1942 

0.1959 

0.1977 

0.1994 

0.2011 

0.2028 

0.2045 

0.2062 

11 

cos 

0.9816 

0.9813 

0.9810 

0.9806 

0.9803 

0.9799 

0.9796 

0.9792 

0.9789 

0.9785 

tan 

0.1944 

0.1962 

0.1980 

0.1998 

0.2016 

0.2035 

0.2053 

0.2071 

0.2089 

0.2107 

sin 

0.2079 

02096 

0.2113 

0.2130 

0.2147 

0.2164 

0.2181 

0.2198 

0.2215 

0.2232 

12 

cos 

0.9781 

0.9778 

0.9774 

0.9770 

0.9767 

0.9763 

0.9759 

0.9755 

0.9751 

0.9748 

tan 

0.2126 

0.2144 

0.2162 

0.2180 

0.2199 

0.2217 

0.2235 

0.2254 

0.2272 

0.2290 

sin 

0.2250 

0.2267 

0.2284 

0.2300 

0.2317 

0.2334 

0.2351 

0.2368 

0.2385 

0.2402 

13 

cos 

0.9744 

0.9740 

0.9736 

9.9732 

0.9728 

0.9724 

0.9720 

0.9715 

0.9711 

0.9707 

tan 

0.2309 

0.2327 

0.2345 

0.2364 

0.2382 

0.2401 

0.2419 

0.2438 

0.2456 

0.2475 

sin 

0.2419 

0.2436 

0.2453 

0.2470 

0.2487 

0.2504 

0.2521 

0.2538 

0.2554 

0.2571 

14 

cos 

0.9703 

0.9699 

0.9694 

0.9690 

0.9686 

0.9681 

0.9677 

0.9673 

0.9668 

0.9664 

tan 

0.2493 

0.2512 

0.2530 

0.2549 

0.2568 

0.2586 

0.2605 

0.2623 

0.2642 

0.2661 

sin 

0.2588 

0.2605 

0.2622 

0.2639 

0.2656 

0.2672 

0.2689 

0.2706 

0.2723 

0.2740 

15 

cos 

0.9659 

0.9655 

0.9650 

0.9646 

0.9641 

0.9636 

0.9632 

0.9627 

0.9622 

0.9617 

tan 

0.2679 

0.2698 

0.2717 

0.2736 

0.2754 

0.2773 

0.2792 

0.2811 

0.2830 

0.2849 

1-22 


MATHEMATICS,   UNITS,   AND   SYMBOLS 


Trigonometric  Functions 


16.0°-35.9° 


Angle 
in 
Degrees 

Name 
of 
Function 

Value  of  Function  for  Each  Tenth  of  a  Degree 

0.0 

0.1 

0.2 

0.3 

0.4 

0.5 

0.6 

0.7 

0.8 

0.9 

—  

sin 

0.2756 

0.2773 

0.2790 

0.2807 

0.2823 

0  2840 

0.2857 

0.2874 

0.2890 

0.2907 

16 

cos 

0.9613 

0.9608 

0.9603 

0.9598 

0.9593 

0.9588 

0.9583 

0.9578 

0.9573 

0.9568 

tan 

0.2867 

0.2886 

0.2905 

0.2924 

0.2943 

0.2962 

0.2981 

0.3000 

0.3019 

0.3038 

sin 

0.2924 

0.2940 

0.2957 

0.2974 

0.2990 

0.3007 

0.3024 

0.3040 

0.3057 

0.3074 

17 

cos 

0.9563 

0.9558 

0.9553 

0.9548 

0  9542 

0.9537 

0.9532 

0.9527 

0.9521 

0.9516 

tan 

0.3057 

0.3076 

0.3096 

0.3115 

0.3134 

0.3153 

0.3172 

0.3191 

0.3211 

0.3230 

sin 

0.3090 

0.3107 

0.3123 

0.3140 

0.3156 

0.3173 

0.3190 

0.3206 

0.3223 

0.3239 

18 

cos 

0.95II 

0.9505 

0.9500 

0.9494 

0.9489 

0.9483 

0.9478 

0.9472 

0.9466 

0.9461 

tan 

0.3249 

0.3269 

0.3288 

0.3307 

0.3327 

0.3346 

0.3365 

0.3385 

0.3404 

0.3424 

sin 

0.3256 

0.3272 

0.3289 

0.3305 

0.3322 

0.3338 

0.3355 

0.3371 

0.3387 

0.3404 

19 

cos 

0  9455 

0.9449 

0.9444 

0.9438 

0.9432 

0.9426 

0.9421 

0.9415 

0.9409 

0.9403 

tan 

0.3443 

0.3463 

0.3482 

0.3502 

0.3522 

0.3541 

0.3561 

0.3581 

0.3600 

0.3620 

sin 

0.3420 

0.3437 

0.3453 

0.3469 

0.3486 

0.3502 

0.3518 

0.3535 

0.3551 

0.3567 

20 

cos 

0.9397 

0.9391 

0.9385 

0.9379 

0.9373 

0.9367 

0.9361 

0.9354 

0.9348 

0.9342 

tan 

0.3640 

0.3659 

0.3679 

0.3699 

0.3719 

0.3739 

0.3759 

0.3779 

0.3799 

0.3819 

sin 

0.3584 

0.3600 

0.3616 

0.3633 

0.3649 

0.3665 

0.3681 

0.3697 

0.3714 

0.3730 

21 

cos 

0.9336 

0.9330 

0.9323 

0.9317 

0.9311 

0.9304 

0.9298 

0.9291 

0.9285 

0.9278 

tan 

0.3839 

0.3859 

0.3879 

0.3899 

0.3919 

0.3939 

0.3959 

0.3979 

0.4000 

0.4020 

sin 

0.3746 

0.3762 

0.3778 

0.3795 

0.3811 

0.3827 

0.3843 

0.3859 

0.3875 

0.3891 

22 

cos 

0.9272 

0.9265 

0.9259 

0.9252 

0.9245 

0.9239 

0.9232 

0.9225 

0.9219 

0.9212 

tan 

0.4040 

0.4061 

0.4081 

0.4101 

0.4122 

0.4142 

0.4163 

0.4183 

0.4204 

0.4224 

sin 

0.3907 

0.3923 

0.3939 

0.3955 

0.3971 

0.3987 

0.4003 

0.4019 

0.4035 

0.4051 

23 

cos 

0.9205 

0.9198 

0.9191 

0.9184 

0.9178 

0.9171 

0.9164 

0.9157 

0.9150 

0.9143 

tan 

0.4245 

0.4265 

0.4286 

0.4307 

0.4327 

0.4348 

0.4369 

0.4390 

0.4411 

0.4431 

sin 

0.4067 

0.4083 

0.4099 

0.4115 

0.4131 

0.4147 

0.4163 

0.4179 

0.4195 

0.4210 

24 

cos 

0.9135 

0.9128 

0.9121 

0.9114 

0.9107 

0.9100 

0.9092 

0.9085 

0.9078 

0.9070 

tan 

0.4452 

0.4473 

0.4494 

0.4515 

0.4536 

0.4557 

0.4578 

0.4599 

0.4621 

0.4642 

sin 

0.4226 

0.4242 

0.4258 

0.4274 

0.4289 

0.4305 

0.4321 

0.4337 

0.4352 

0.4368 

25 

cos 

0.9063 

0.9056 

0.9048 

0.9041 

0.9033 

0.9026 

0.9018 

0.9011 

0.9003 

0.8996 

tan 

0.4663 

0.4684 

0.4706 

0.4727 

0.4748 

0.4770 

0.4791 

0.4813 

0.4834 

0.4856 

sin 

0.4384 

0.4399 

0.4415 

0.4431 

0.4446 

0.4462 

0.4478 

0.4493 

0.4509 

0.4524 

26 

cos 

0.8988 

0.8980 

0.8973 

0.8965 

0.8957 

0.8949 

0.8942 

0.8934 

0.8926 

0.8918 

tan 

0.4877 

0.4899 

0.4921 

0.4942 

0.4964 

0.4986 

0.5008 

0.5029 

0.5051 

0.5073 

sin 

0.4540 

0.4555 

0.4571 

0.4586 

0.4602 

0.4617 

0.4633 

0.4648 

0.4664 

0.4679 

27 

cos 

0.8910 

0.8902 

0.8894 

0.8886 

0.8878 

0.8870 

0.8862 

0.8854 

0.8846 

0.8838 

tan 

0.5095 

0.5117 

0.5139 

0.5161 

0.5184 

0.5206 

0.5228 

0.5250 

0.5272 

0.5295 

sin 

0.4695 

0.4710 

0.4726 

0.4741 

0.4756 

0.4772 

0.4787 

0.4802 

0.4818 

0.4833 

28 

cos 

0.8829 

0.8821 

0.8813 

0.8805 

0.8796 

0.8788 

0.8780 

0.8771 

0.8763 

0.8755 

tan 

0.53J7 

0.5340 

0.5362 

0.5384 

0.5407 

0.5430 

0.5452 

0.5475 

0.5498 

0.5520 

sin 

0.4848 

0.4863 

0.4879 

0.4894 

0.4909 

0.4924 

0.4939 

0.4955 

0.4970 

0.4985 

29 

cos 

0.8746 

0.8738 

0.8729 

0.8721 

0.8712 

0.8704 

0.8695 

0.8686 

0.8678 

0.8669 

tan 

0.5543 

0,5566 

0.5589 

0.5612 

0.5635 

0.5658 

0.5681 

0.5704 

0.5727 

0.5750 

sin 

0.5000 

0.5015 

0.5030 

0.5045 

0.5060 

0.5075 

0.5090 

0.5105 

0.5120 

0.5135 

30 

cos 

0.8660 

0.8652 

0.8643 

0.8634 

0.8625 

0.8616 

0  8607 

0.8599 

0.8590 

0.8581 

tan 

0.5774 

0.5797 

0.5820 

0.5844 

0.5867 

0.5890 

0.5914 

0.5938 

0.5961 

0.5985 

sin 

0.5150 

0.5165 

0.5180 

0.5195 

0.5210 

0.5225 

0.5240 

0.5255 

0.5270 

0.5284 

31 

cos 

0.8572 

0.8563 

0.8554 

0.8545 

0.8536 

0.8526 

0.8517 

0.8508 

0.8499 

0.8490 

fofl 

0.6009 

0.6032 

0.6056 

0.6080 

0,6104 

0.6128 

0.6152 

0.6176 

0.6200 

0.6224 

sin 

0.5299 

0.5314 

0.5329 

0.5344 

0.5358 

0.5373 

0.5388 

0.5402 

0.5417 

0.5432 

32 

cos 

0.8480 

0.8471 

0.8462 

0.8453 

0.8443 

0.8434 

0.8425 

0.8415 

0.8406 

0.8396 

tan 

0.6249 

0.6273 

0.6297 

0.6322 

0.6346 

0.6371 

0.6395 

0.6420 

0.6445 

0.6469 

sin 

0.5446 

0.5461 

0.5476 

0.5490 

0.5505 

0.5519 

0.5534 

0.5548 

0.5563 

0.5577 

33 

cos 

0.8387 

0.8377 

0.8368 

0.8358 

0.8348 

0.8339 

0.8329 

0.8320 

0.8310 

0.8300 

tan 

0.6494 

0.6519 

0.6544 

0.6569 

0.6594 

0.6619 

0.6644 

0.6669 

0.6694 

0.6720 

sin 

0.5592 

0.5606 

0.5621 

0.5635 

0.5650 

0.5664 

0.5678 

0.5693 

0.5707 

0.5721 

34 

cos 

0.8290 

0.8281 

0.8271 

0.8261 

0.8251 

0.8241 

0.8231 

0.8221 

0.8211 

0.8202 

tan 

0.6745 

0.6771 

0.6796 

0.6822 

0.6847 

0.6873 

0.6899 

0.6924 

0.6950 

0.6976 

sin 

0.5736 

0.5750 

0.5764 

0.5779 

0.5793 

0.5807 

0.5821 

0.5835 

0.5850 

0.5864 

35 

cos 

0.8192 

0.8181 

0.8I7I 

0.8161 

0.8151 

0.8141 

0,8131 

0,8121 

0.8111 

0.8100 

tan 

0.7002 

0.7028 

0.7054 

0.7080 

0.7107 

0,7133 

0,7159 

0,7186 

0.7212 

0.7239 

TRIGONOMETRIC  TABLES 


1-23 


Trigonometric  Functions 


36.0°-55.9° 


Angle 
in 
Degrees 

Name 
of 
Function 

Value  of  Function  for  Each  Tenth  of  a  Degree 

0.0 

0.1 

0.2 

0.3 

0.4 

0.5 

0.6 

0.7 

0.8 

0.9 

sin 

0.5878 

0.5892 

0.5906 

0.5920 

0.5934 

0.5948 

0.5962 

0.5976 

0.5990 

0.6004 

36 

cos 

0.8090 

0.8080 

0.8070 

0.8059 

0.8049 

0.8039 

0.8028 

0.8018 

0.8007 

0.7997 

tan 

0.7265 

0.7292 

0.7319 

0.7346 

0.7373 

0.7400 

0.7427 

0.7454 

0.7481 

0.7508 

sin 

0.6018 

0.6032 

0.6046 

0.6060 

0.6074 

0.6088 

0.6101 

0.6115 

0.6129 

0.6143 

37 

cos 

0.7986 

0.7976 

0  7965 

0.7955 

0.7944 

0.7934 

0.7923 

0.7912 

0.7902 

0.7891 

tan 

0.7536 

0.7563 

0.7590 

0.7618 

0.7646 

0.7673 

0.7701 

0.7729 

0.7757 

0.7785 

sin 

0.6157 

0.6170 

0.6184 

0.6198 

0.6211 

0.6225 

0.6239 

0.6252 

0.6266 

0.6280 

38 

cos 

0.7880 

0.7869 

0.7859 

0.7848 

0.7837 

0.7826 

0.7815 

0.7804 

0.7793 

0.7782 

tan 

0.7813 

0.7841 

0.7869 

0.7898 

0.7926 

0.7954 

0.7983 

0.8012 

0.8040 

0.8069 

sin 

0.6293 

0.6307 

0.6320 

0.6334 

0.6347 

0.6361 

0.6374 

0.6388 

0.6401 

0.6414 

39 

cos 

0.7771 

0.7760 

0.7749 

0.7738 

0.7727 

0.7716 

0.7705 

0.7694 

0.7683 

0.7672 

tan 

0.8098 

0.8127 

0.8156 

0.8185 

0.8214 

0.8243 

0.8273 

0.8302 

0.8332 

0.8361 

sin 

0.6428 

0.6441 

0,6455 

0.6468 

0.6481 

0.6494 

0.6508 

0.6521 

0.6534 

0.6547 

40 

cos 

0.7660 

0.7649 

0.7638 

0.7627 

0.7615 

0.7604 

0.7593 

0.7581 

0.7570 

0.7559 

tan 

0.8391 

0.8421 

0.8451 

0.8481 

0.8511 

0.8541 

0.8571 

0.8601 

0.8632 

0.8662 

sin 

0.6561 

0.6574 

0.6587 

0.6600 

0.6613 

0.6626 

0.6639 

0.6653 

0.6665 

0.6678 

41 

cos 

0.7547 

0.7536 

0.7524 

0.7513 

0.7501 

0.7490 

0.7478 

0.7466 

0.7455 

0.7443 

tan 

0.8693 

0.8724 

0.8754 

0.8785 

0.8816 

0.8847 

0.8878 

0.8910 

0.8941 

0.8972 

sin 

0.6691 

0.6704 

0.6717 

0.6730 

0.6743 

0.6756 

0.6769 

0.6782 

0.6794 

0.6807 

42 

cos 

0.7431 

0.7420 

0.7408 

0.7396 

0.7385 

0.7373 

0.7361 

0.7349 

0.7337 

0.7325 

tan 

0.9004 

0.9036 

0.9067 

0.9099 

0.9131 

0.9163 

0.9195 

0.9228 

0.9260 

0.9293 

sin 

0.6820 

0.6833 

0.6845 

0.6858 

0.6871 

0.6884 

0.6896 

0.6909 

0.6921 

0.6934 

43 

cos 

0.7314 

0.7302 

0.7290 

0.7278 

0.7266 

0.7254 

0.7242 

0.7230 

0.7218 

0.7206 

tan 

0.9325 

0.9358 

0.9391 

0.9424 

0.9457 

0.9490 

0.9523 

0.9556 

0.9590 

0.9623 

sin 

0.6947 

0.6959 

0.6972 

0.6984 

0.6997 

0.7009 

0.7022 

0.7034 

0.7046 

0.7059 

44 

cos 

0.7193 

0.7181 

0.7169 

0.7157 

0.7145 

0.7133 

0.7120 

0.7108 

0.7096 

0.7083 

tan 

0.9657 

0.9691 

0.9725 

0.9759 

0.9793 

0.9827 

0.9861 

0.9896 

0.9930 

0.9965 

sin 

0.7071 

0.7083 

0.7096 

0.7108 

0.7120 

0.7133 

0.7145 

0.7157 

0.7169 

0.7181 

45 

coa 

0.7071 

0.7059 

0.7046 

0.7034 

0.7022 

0.7009 

0.6997 

0.6984 

0.6972 

0.6959 

tan 

I.  0000 

1.0035 

1.0070 

1.0105 

1.0141 

1.0176 

1.0212 

1.0247 

1.0283 

1.0319 

sin 

0.7193 

0.7206 

0.7218 

0.7230 

0.7242 

0.7254 

0.7266 

0.7278 

0.7290 

0.7302 

46 

cos 

0.6947 

0.6934 

0.6921 

0.6909 

0.6896 

0.6884 

0.6871 

0.6858 

0.6845 

0.6833 

tan 

1.0355 

1.0392 

1.0428 

1.0464 

1.0501 

1.0538 

1.0575 

1.0612 

1.0649 

1.0686 

sin 

0.7314 

0.7325 

0.7337 

0.7349 

0.7361 

0.7373 

0.7385 

0.7396 

0.7408 

0.7420 

47 

cos 

0.6820 

0.6807 

0.6794 

0.6782 

0.6769 

0.6756 

0.6743 

0.6730 

0.6717 

0.6704 

tan 

1.0724 

1.0761 

1.0799 

1.0837 

1.0875 

1.0913 

1.0951 

I  .0990 

1.1028 

1.1067 

sin 

0.7431 

0.7443 

0.7455 

0.7466 

0.7478 

0.7490 

0.7501 

0.7513 

0.7524 

0.7536 

48 

cos 

0.6691 

0.6678 

0.6665 

0.6652 

0.6639 

0.6626 

0.6613 

0.6600 

0.6587 

0.6574 

tan 

1.1106 

1.1145 

1.1184 

1.1224 

1.1263 

1.1303 

1.1343 

1.1383 

1.1423 

1.1463 

sin 

0.7547 

0.7559 

0.7570 

0.7581 

0.7593 

0.7604 

0.7615 

0.7627 

0.7638 

0.7649 

49 

cos 

0.6561 

0.6547 

0.6534 

0.6521 

0.6508 

0.6494 

0.6481 

0.6468 

0.6455 

0.6441 

tan 

1  .  1504 

1.1544 

1.1585 

1.1626 

1.1667 

1.1708 

1.1750 

1.1792 

1.1833 

1.1875 

sin 

0.7660 

0.7672 

0.7683 

0.7694 

0.7705 

0.7716 

0.7727 

0.7738 

0.7749 

0.7760 

50 

cos 

0.6428 

0.6414 

0.6401 

0.6388 

0.6374 

0.6361 

0.6347 

0.6334 

0.6320 

0.6307 

tan 

1.1918 

1.1960 

1.2002 

1  .2045 

1  .2088 

1.2131 

1.2174 

1.2218 

1.2261 

1.2305 

sin 

0.7771 

0.7782 

0.7793 

0.7804 

0.7815 

0.7826 

0.7837 

0.7848 

0.7859 

0.7869 

51 

cos 

0.6293 

0.6280 

0.6266 

0.6252 

0.6239 

0.6225 

0.6211 

0.6198 

0.6184 

0.6170 

tan 

1.2349 

1  .2393 

1  .2437 

1  .2482 

1.2527 

1.2572 

1.2617 

1.2662 

I.  2708 

1.2753 

sin 

0.7880 

0.789.1 

0.7902 

0.7912 

0.7923 

0.7934 

0.7944 

0.7955 

0.7965 

0.7976 

52 

cos 

0.6157 

0.6143 

0.6129 

0.6115 

0.6101 

0.6088 

0.6074 

0.6060 

0.6046 

0.6032 

tan 

1.2799 

1  .2846 

1  .2892 

1.2938 

1.2985 

1  .3032 

1.3079 

1.3127 

1.3175 

1.3222 

sin 

0.7986 

0.7997 

0.8007 

0.8018 

0.8028 

0.8039 

0.8049 

0.8059 

0.807d 

0.8080 

53 

cos 

0.6018 

0.6004 

0.5990 

0.5976 

0.5962 

0.5948 

0.5934 

0.5920 

0.5906 

0.5892 

tan 

1.3270 

1.3319 

1.3367 

1.3416 

1.3465 

1.3514 

1.3564 

1.3613 

1.3663 

1.3713 

sin 

0.8090 

0.8100 

0.8111 

0.8121 

0.8131 

0.8141 

0.8151 

0.8161 

0.8171 

0.8181 

54 

cos 

0.5878 

0.5864 

0.5850 

0.5835 

0.5821 

0.5807 

0.5793 

0.5779 

0.5764 

0.5750 

tan 

1.3764 

1.3814 

1.3865 

1.3916 

1.3968 

1.4019 

1.4071 

1.4124 

1.4176 

1.4229 

sin 

0.8192 

0.8202 

0.8211 

0.8221 

0.8231 

0.8241 

0.8251 

0.8261 

0.8271 

0.8281 

55 

cos 

0.5736 

0.5721 

0.5707 

0.5693 

0.5678 

0.5664 

0.5650 

0.5635 

0.5621 

0.5606 

tan 

1.4281 

1.4335 

1.4388 

1.4442 

1.4496 

1.4550 

1.4605 

1.4659 

1.4715 

1.4770 

1-24 


MATHEMATICS,  UNITS,  AND  SYMBOLS 


Trigonometric  Functions 


56.0°-75.9° 


Angle 

"~ 
Name 

.  .  —  ~  
Value  of  Function  for  Each  Tenth  of  a  Degree 

in 
Degrees 

of 
Function 

0.0 

0.1 

0.2 

0.3 

0.4 

0.5 

0.6 

0.7 

0.8 

0.9 

56 

sin 
cos 
tan 

.8290 
.5592 
.4826 

0.8300 
0.5577 
1.4882 

0.8310 
0.5563 
1  .4938 

0.8320 
0.5548 
1.4994 

0.8329 
0.5534 
1.5051 

0.8339 
0.5519 
1.5108 

0.8348 
0.5505 
1.5166 

0.8358 
0.5490 
1.5224 

0.8368 
0.5476 
1.5282 

0.8377 
0.5461 
1.5340 

57 

sin 
cos 
tan 

.8387 
.5446 
.5399 

0.8396 
0.5432 
1.5458 

0.8406 
0.5417 
1.5517 

0.8415 
0.5402 
1.5577 

0.8425 
0.5388 
1  .5637 

0.8434 
0.5373 
1.5697 

0.8443 
0.5358 
1.5757 

0.8453 
0.5344 
1.5818 

0.8462 
0.5329 
1.5880 

0.8471 
0.5314 
I  .5941 

58 

sin 
cos 
tan 

0.8480 
0.5299 
.6003 

0.8490 
0.5284 
1.6066 

0.8499 
0.5270 
1.6128 

0.8508 
0.5255 
1.6191 

0.8517 
0.5240 
1.6255 

0.8526 
0.5225 
1.6319 

0.8536 
0.5210 
1.6383 

0.8545 
0.5195 
1.6447 

0.8554 
0.5180 
1.6512 

0.856? 
0.5165 
1.6577 

59 

sin 
cos 
tan 

0.8572 
0.5150 
.6643 

0.8581 
0.5135 
1  .6709 

0.8590 
0.5120 
1.6775 

0.8599 
0.5105 
1.6842 

0.8607 
0.5090 
1  .6909 

0.8616 
0.5075 
1.6977 

0.8625 
0.5060 
1.7045 

0.8634 
0.5045 
1.7113 

0.8643 
0.5030 
1.7182 

0.8652 
0.5015 
1.7251 

60 

sin 
cos 
tan 

0.8660 
0.5000 
.7321 

0.8669 
0.4985 
1.7391 

0.8678 
0.4970 
1.7461 

0.8686 
0.4955 
1.7532 

0.8695 
0.4939 
1.7603 

0.8704 
0.4924 
1.7675 

0.8712 
0.4909 
1.7747 

0.8721 
0.4894 
1.7820 

0.8729 
0.4879 
1.7893 

0.8738 
0.4863 
1.7966 

61 

sin 
cos 

tan 

0.8746 
0.4848 
1.8040 

0.8755 
0.4833 
1.8115 

0.8763 
0.4818 
1.8190 

0.8771 
0.4802 
1.8265 

0.8780 
0.4787 
1.8341 

0.8788 
0.4772 
1.8418 

0.8796 
0.4756 
1.8495 

0.8805 
0.4741 
1.8572 

0.8813 
0.4726 
1.8650 

0.8821 
0.4710 
1.8728 

62 

sin 
cos 
tan 

0.8829 
0.4695 
1.8807 

0.8838 
0.4679 
1.8887 

0.8846 
0.4664 
1.8967 

0.8854 
0.4648 
1.9047 

0.8862 
0.4633 
1.9128 

0.8870 
0.4617 
1.9210 

0.8878 
0.4602 
1  .9292 

0.8886 
0.4586 
1  .9375 

0.8894 
0.4571 
1.9458 

0.8902 
0.4555 
1.9542 

0.8910 

0.8918 

0.8926 

0.8934 

0.8942 

0.8949 

0.8957 

0.8965 

0.8973 

0.8980 

63 

cos 
tan 

0.4540 
1.9626 

0.4524 
1.9711 

0.4509 
1  .9797 

0.4493 
1.9883 

0.4478 
1.9970 

0.4462 
2.0057 

0.4446 
2.0145 

0.4431 
2.0233 

0.4415 
2.0323 

0.4399 
2.0413 

0.8988 

0.8996 

0.9003 

0.9011 

0.9018 

0.9026 

0.9033 

0.9041 

0.9048 

0.9056 

64 

cos 

0.4384 

0.4368 

0.4352 

0.4337 

0.4321 

0.4305 

0.4289 

0.4274 

0.4258 

0.4242 

tan 

2.0503 

2.0594 

2.0686 

2.0778 

2.0872 

2.0965 

2.1060 

2.1155 

2.1251 

2.1348 

. 

0.9063 

0.9070 

0.9078 

0.9085 

0.9092 

0.9100 

0.9107 

0.9114 

0.9121 

0.9128 

65 

cos 

0.4226 

0.4210 

0.4195 

0.4179 

0.4163 

0.4147 

0.4131 

0.4115 

0.4099 

0.4083 

tan 

2.1445 

2.1543 

2.1642 

2.1742 

2.1842 

2.1943 

2.2045 

2.2148 

2.2251 

2.2355 

. 

0.9135 

0.9143 

0.9150 

0.9157 

0.9164 

0.9171 

0.9178 

0.9184 

0.9191 

0.9198 

66 

cos 

0.4067 

0.4051 

0.4035 

0.4019 

0.4003 

0.3987 

0.3971 

0.3955 

0.3939 

0.3923 

tan 

2.2460 

2.2566 

2.2673 

2.2781 

2.2889 

2.2998 

2.3109 

2.3220 

2.3332 

2.3445 

sin 

0.9205 

0.9212 

0.9219 

0.9225 

0.9232 

0.9239 

0.9245 

0.9252 

0.9259 

0.9265 

67 

cos 

0.3907 

0.3891 

0.3875 

0,3859 

0.3843 

0.3827 

0.3811 

0.3795 

0.3778 

0.3762 

tan 

2.3559 

2.3673 

2.3789 

2.3906 

2.4023 

2.4142 

2.4262 

2.4383 

2.4504 

2.4627 

sin 

0.9272 

0.9278 

0.9285 

0.9291 

0.9298 

0.9304 

0.9311 

0.9317 

0.9323 

0.9330 

68 

cos 

0.3746 

0.3730 

0.3714 

0.3697 

0.3681 

0.3665 

0.3649 

0.3633 

0.3616 

0.3600 

tan 

2.4751 

2.4876 

2.5002 

2.5129 

2.5257 

2.5386 

2.5517 

2.5649 

2.5782 

2.5916 

sin 

0.9336 

0.9342 

0.9348 

0.9354 

0.9361 

0.9367 

0.9373 

0.9379 

0.9385 

0.9391 

69 

cos 

0.3584 

0.3567 

0.3551 

0.3535 

0.3518 

0.3502 

0.3486 

0.3469 

0.3453 

0.3437 

tan 

2.6051 

2.6187 

2.6325 

2.6464 

2.6605 

2.6746 

2.6889 

2.7034 

2.7179 

2.7326 

sin 

0.9397 

0.9403 

0.9409 

0.9415 

0.9421 

0.9426 

0.9432 

0.9438 

0.9444 

0.9449 

70 

cos 

0.3420 

0.3404 

0,3387 

0.3371 

0.3355 

0.3338 

0.3322 

0.3305 

0.3289 

0.3272 

tan 

2.7475 

2.7625 

2.7776 

2.7929 

2.8083 

2.8239 

2.8397 

2.8556 

2.8716 

2.8878 

sin 

0.9455 

0.9461 

0.9466 

0.9472 

0.9478 

0.9483 

0.9489 

0.9494 

0.9500 

0.9505 

71 

cos 

0.3256 

0.3239 

0.3223 

0.3206 

0.3190 

0.3173 

0.3156 

0.3140 

0.3123 

0.3107 

tan 

2.9042 

2.9208 

2.9375 

2.9544 

2.9714 

2.9887 

3.0061 

3.0237 

3.0415 

3.0595 

sin 

0.9511 

0.9516 

0.9521 

0.9527 

0.9532 

0.9537. 

0.9542 

0.9548 

0.9553 

0.9558 

72 

cos 

0.3090 

0.3074 

0.3057 

0.3040 

0.3024 

0.3007 

0.2990 

0.2974 

0.2957 

0.2940 

tan 

3.0777 

3.0961 

3.1146 

3.1334 

3.1524 

3.1716 

3.1910 

3.2106 

3.2305 

3.2506 

sin. 

0.9563 

0.9568 

0.9573 

0.9578 

0.9583 

0.9588 

0.9593 

0.9598 

0.9603 

0.9608 

73 

COB 

0.2924 

0.2907 

0.2890 

0.2874 

0.2857 

0.2840 

0.2823 

0.2807 

0.2790 

0.2773 

tan 

3.2709 

3.2914 

3.3122 

3.3332 

3.3544 

3.3759 

3.3977 

3.4197 

3.4420 

3.4646 

sin 

0.9613 

0.9617 

0.9622 

0.9627 

0.9632 

0.9636 

0.9641 

0.9646 

0.9650 

0.9655 

74 

cos 

0.2756 

0.2740 

0.2723 

0.2706 

0.2689 

0.2672 

0.2656 

0.2639 

0.2622 

0.2605 

tan 

3.4874 

3.5105 

3.5339 

3.5576 

3.5816 

3.6059 

3.6305 

3.6554 

3.6806 

3.7062 

sin 

0.9659 

0.9664 

0.9668 

0.9673 

0.9677 

0.9681 

0.9686 

0.9690 

0.9694 

0.9699 

75 

COB 

0.2588 

0.2571 

0.2554 

0.2538 

0.2521 

0.2504 

0.2487 

0.2470 

0.2453 

0.2436 

tan 

3.7321 

3.7583 

3.7848 

3.8118 

3.8391 

3.8667 

3.8947 

3.9232 

3.9520 

3.9812 

TRIGONOMETRIC  TABLES 


1-25 


Trigonometric  Functions 


76.0°-89.9° 


Angle 
in 
Degrees 

Name 
of 
Function 

Value  of  Function  for  Each  Tenth  of  a  Degree 

0.0 

0.1 

0.2 

0.3 

0.4 

0.5 

0.6 

0.7 

0.8 

0.9 

76 

sin 
cos 
tan 

0.9703 
0.2419 
4.0108 

0.9707 
0.2402 
4.0408 

0.9711 
0.2385 
4.0713 

0.9715 
0.2368 
4.1022 

0.9720 
0.2351 
4.1335 

0.9724 
0.2334 
4.1653 

0.9728 
0.2317 
4.1976 

0.9732 
0.2300 
4.2303 

0.9736 
0.2284 
4.2635 

0.9740 
0.2267 
4.2972 

77 

sin 
cos 
tan 

0.9744 
0.2250 
4.3315 

0.9748 
0.2232 
4.3662 

0.9751 
0.2215 
4.4015 

0.9755 
0.2198 
4.4374 

0.9759 
0.2181 

4.4737 

0.9763 
0.2164 
4.5107 

0.9767 
0.2147 
4.5483 

0.9770 
0.2130 
4.5864 

0.9774 
0.2113 
4.6252 

0.9778 
0.2096 
4.6646 

78 

sin 
cos 
tan 

0.9781 
0.2079 
4.7046 

0.9785 
0.2062 
4.7453 

0.9789 
0.2045 
4.7867 

0.9792 
0.2028 
4.8288 

0.9796 
0.2011 
4.8716 

0.9799 
0.1994 
4.9152 

0.9803 
0.1977 
4.9594 

0.9806 
0.1959 
5.0045 

0.9810 
0.1942 
5.0504 

0.9813 
0.1925 
5.0970 

79 

sin 
cos 
tan 

0.9816 
0.1908 
5.1446 

0.9820 
0.1891 
5.1929 

0.9823 
0.1874 
5.2422 

0.9826 
0.1857 
5.2924 

0.9829 
0.1840 
5.3435 

0.9833 
0.1822 
5.3955 

0.9836 
0.1805 
5.4486 

0.9839 
0.1788 
5.5026 

0.9842 
0.1771 
5.5578 

0.9845 
0.1754 
5.6140 

80 

sin 
cos 
tan 

0.9848 
0.1736 
5.6713 

0.9851 
0.1719 
5.7297 

0.9854 
0.1702 
5.7894 

0.9857 
0.1685 
5.8502 

0.9860 
0.1668 
5.9124 

0.9863 
0.1650 
5.9758 

0.9866 
0.1633 
6.0405 

0.9869 
0.1616 
6.1066 

0.9871 
0.1599 
6.1742 

0.9874 
0.1582 
6.2432 

81 

sin 
cos 
tan 

0.9877 
0.1564 
6.3138 

0.9880 
0.1547 
6.3859 

0.9882 
0.1530 
6.4596 

0.9885 
0.1513 
6.5350 

0.9888 
0.1495 
6.6122 

0.9890 
0.1478 
6.6912 

0.9893 
0.1461 
6.7720 

0.9895 
0.1444 
6.8548 

0.9898 
0.1426 
6.9395 

0.9900 
0.1409 
7.0264 

82 

sin 
cos 
tan 

0.9903 
0.1392 
7.1154 

0.9905 
0.1374 
7.2066 

0.9907 
0.1357 
7.3002 

0.9910 
0.1340 
7.3962 

0.9912 
0.1323 
7.4947 

0.9914 
0.1305 
7.5958 

0.9917 
0.1288 
7.6996 

0.9919 
0.1271 
7.8062 

0.9921 
0.1253 
7.9158 

0.9923 
0.1236 
8.0285 

83 

sin 
cos 
tan 

0.9925 
0.1219 
8.1443 

0.9928 
0.1201 
8.2636 

0.9930 
0.1184 
8.3863 

0.9932 
0.1167 
8.5126 

0.9934 
0.1149 
8.6427 

0.9936 
0.1132 
8.7769 

0.9938 
0.1115 
8.9152 

0.9940 
0.1097 
9.0579 

0.9942 
0.1080 
9.2052 

0.9943 
0.1063 
9.3572 

84 

sin 
cos 
tan 

0.9945 
0.1045 
9.5144 

0.9947 
0.1028 
9.6768 

0.9949 
0.1011 
9.8448 

0.9951 
0.0993 
10.02 

0.9952 
0.0976 
10.20 

0.9954 
0.0958 
10.39 

0.9956 
0.0941 
10.58 

0.9957 
0.0924 
10.78 

0.9959 
0.0906 
10.99 

0.9960 
0.0889 
11.20 

85 

sin 
cos 
tan 

0.9962 
0.0872 
11.43 

0.9963 
0.0854 
11.66 

0.9965 
0.0837 
11.91 

0.9966 
0.0819 
12.16 

0.9968 
0.0802 
12.43 

0.9969 
0.0785 
12.71 

0.9971 
0.0767 
13.00 

0.9972 
0.0750 
13.30 

0.9973 
0.0732 
13.62 

0.9974 
0.0715 
13.95 

86 

sin 
cos 
tan 

0.9976 
0.0698 
14.30 

0.9977 
0.0680 
14.67 

0.9978 
0.0663 
15.06 

0.9979 
0.0645 
15.89 

0.9980 
0.0628 
15.46 

0.9981 
0.0610 
16.35 

0.9982 
0.0593 
16.83 

0.9983 
0.0576 
17.34 

0.9984 
0.0558 
17.89 

0.9985 
0.0541 
18.46 

87 

sin 
cos 
tan 

0.9986 
0.0523 
19.08 

0.9987 
0.0506 
19.74 

0.9988 
0.0488 
20.45 

0.9989 
0.0471 
21.20 

0.9990 
0.0454 
22.02 

0.9990 
0.0436 
22.90 

0.9991 
0.0419 
23.86 

0,9992 
0.0401 
24.90 

0.9993 
0.0384 
26.03 

0.9993 
0.0366 
27.27 

88 

sin 
cos 
tan 

0.9994 
0.0349 
28.64 

0.9995 
0.0332 
30.14 

0.9995 
0.0314 
31.82 

0.9996 
0.0297 
33.69 

0.9996 
0.0279 
35.80 

0.9997 
0.0262 
38.19 

0.9997 
0.0244 
40.92 

0.9997 
0.0227 
44.07 

0.9998 
0.0209 
47.74 

0.9998 
0.0192 
52.08 

89 

sin 
cos 
tan 

0.9998 
0.0175 
57.29 

0.9999 
0.0157 
63.66 

0.9999 
0.0140 
71.62 

0.9999 
,0.0122 
81.85 

0.9999 
0.0105 
95.49 

1.000 
0.0087 
114.6 

1.000 
0.0070 
H3.2    fe 

1.000 
0.0052 
191.0 

1.000 
0.0035 
286.5 

1.000 
0.0017 
573.0 

1-26 


MATHEMATICS;  UNITS,  AND  SYMBOLS 


13.    EXPONENTIAL  AND  HYPERBOLIC  TABLES 

The  following  tables  give  values  of  «*,  e"*,  sinh  x,  cosh  x  and  tanh  x  for  values  of  a: 
from  0.00  to  6.00  in  intervals  of  0.01. 

To  facilitate  computations  involving  multiplication,  the  common  logarithms  of  ex, 
sinh  xy  cosh  x,  and  tanh  x  are  also  given. 

For  values  of  x  greater  than  6,  e*  may  be  computed  from  the  relationship  eT  =  log"1 
(x  logio  e)  =  log""1  0.43429#;  e~x  approaches  zero;  sinh  x  and  cosh  x  are  approximately 
equal  and  become  0.5  e*;  and  tanh  x  and  coth  x  have  values  approximately  equal  to  unity. 

Where  more  accurate  values  of  the  exponentials  and  functions  are  required  they  may 
be  computed  from  the  following  relationships. 


e  =  2.71828  18285 

M  «  logio  e  =  0.43429  44819 
e*  =  log-1  M:c< 


-  =  0.36787  94412 
e 


—  «  loge  10 
M 


2.30258  50930 
log"1  —  MX 


Values  of  Hyperbolic  Functions  0 

*-«  r\>  «*>  4*  §  p 

Q.X  = 

IX  = 

2 
1    * 

—             ^cosna;  = 
secha;  = 

2 
1 

tanna;  = 
coth  a;  = 

1 

sinh  # 

cosh  x 

tanh  a: 

ft 

7) 

1 

J 

jf 

I 

77 

1 

7 

I 

I 

L 

/ 

'( 

1 

/ 

I 

y 

/ 

I 

y 

/ 

H 

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^  i 

H 

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\\ 

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1 

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^ 

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^ 

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1 

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/ 

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\ 

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ffll 

\ 

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L- 

s 

( 

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s 

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f* 

s^ 

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/ 

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3 

4 

Hyperbolic  Radians  (x) 
Chart  of  the  Hyperbolic  Functions. 


EXPONENTIAL  AND  HYPERBOLIC  TABLES 


1-27 


X 

Natural  Values 

Common  Logarithms 

6s 

e~* 

Sinh.3 

Cosh  a; 

Tanha? 

G* 

Sink  a: 

Coshx 

Tanha; 

0.00 

1.0000 

1.0000 

0.0000 

1.0000 

.00000 

0.00000 

—  00 

0.00000 

—  00 

0.01 
0.02 
0.03 

.0101 
.0202 
.0305 

.99005 
.98020 
.97045 

0.0100 
0.0200 
0.0300 

1.0001 
1.0002 
1.0005 

.01000 
.02000 
.02999 

.00434 
.00869 
.01303 

2.00001 
.30106 
.47719 

.00002 
,00009 
.00020 

3.99999 
2.30097 
.47699 

0.04 
0.05 
0.06 

.0408 
.0513 
.0618 

.  96079 
.95123 
.94176 

0.  0400 
0.0500 
0.0600 

1.0008 
1.0013 
1.0018 

.03998 
.04996 
.05993 

.01737 
.02171 
.02606 

.60218 
.69915 
.77841 

.00035 
.00054 
.00078 

.60183 
.69861 
.77763 

0.07 
0.08 
0.09 

.0725 
.0833 
.0942 

,93239 
.92312 
.91393 

0.0701 
0.0801 
0.0901 

1.0025 
1.0032 
1.0041 

.06989 
.07983 
.08976 

.03040 
.03474 
.03909 

.84545 
.90355 
.  95483 

.00106 
.00139 
.00176 

.84439 
.90216 
.95307 

0.10 

.1052 

.90484 

0.1002 

1.0050 

.09967 

0.04343 

1.00072 

0.00217 

2.99856 

0.11 
0.12 
0.13 

.1163 
.1275 
.1388 

.89583 
.88692 
.87810 

0.1102 
0.1203 
0.1304 

1.0061 
1.0072 
1.0085 

.10956 
.11943 
.12927 

.04777 
.05212 
.05646 

.04227 
.  08022 
.11517 

.00262 
.00312 
.00366 

1.03965 
.07710 
.11151 

0.14 
0.15 
0.  16 

.1503 
.1618 
.1735 

.86936 
.86071 
.85214 

0.1405 
0.1506 
0.1607 

1.0098 
1.0113 
1.0128 

.13909 
.14889 
.15865 

.06080 
,06514 
.  06949 

.  14755 
.17772 
.  20597 

.00424 
.00487 
.00554 

.14330 
.17285 
.  20044 

0.17 
0.18 
0,  19 

.1853 
.1972 
.2092 

.84366 
.83527 
.82696 

0.1708 
0.1810 
0.1911 

1.0145 
1.0162 
1.0181 

.16838 
.17808 
.18775 

.07383 
.07817 
.08252 

.23254 
.25762 
.28136 

.00625 
.00700 
.00779 

.22629 
.25062 
.27357 

0.20 

.2214 

.81873 

0.2013 

1.0201 

.19738 

0.08686 

1.30392 

0.00863 

1.29529 

0.21 
0.22 
0.23 

.2337 
.2461 
.2586 

.81058 
.80252 
.79453 

0.2115 
0.2218 
0.2320 

1.0221 
1.0243 
1.0266 

.20697 
.21652 
.  22603 

.09120 
.09554 
.09989 

.32541 
.34592 
.36555 

.00951 
.01043 
.01139 

.31590 
.33549 
.35416 

0.24 
0.25 
0.26 

.2712 
.2840 
.2969 

.78663 
.77880 
.77105 

0.2423 
0.2526 
0.2629 

1  .  0289 
1.0314 
1.0340 

.23550 
.24492 
.  25430 

.10423 
.10857 
.11292 

.38437 
.40245 
.41986 

.01239 
.01343 
.01452 

.37198 
.38902 
.40534 

0.27 
0.28 
0.29 

.3100 
.3231 
.3364 

.76338 

.75578 
.74826 

0.2733 
0.2837 
0.2941 

1.0367 
1.0395 
1.0423 

.26362 
.27291 
.28213 

.11726 
.12160 
.12595 

.43663 
.45282 
.46847 

.01564 
.01681 
.01801 

.42099 
.43601 
.45046 

0.30 

.3499 

.74082 

0.3045 

1.0453 

.29131 

0.13029 

1.48362 

0.01926 

1.46436 

0.31 
0.32 
0.33 

.3634 
.3771 
.3910 

.73345 
.72615 
.71892 

0.3150 
0.3255 
0.3360 

1.0484 
1.0516 
1  .  0549 

.30044 
.30951 
.31852 

.13463 
.13897 
.14332 

.49830 
.51254 
.52637 

.02054 
.02107 
.02323 

.47775 
.49067 
.50314 

0.34 
0.35 
0.36 

.4049 
.4191 
.4333 

.71177 
.70469 
,69768 

0.3466 
0.3572 
0.3678 

1.0584 
1.0619 
1.0655 

.32748 
.33638 
.34521 

.14766 
.15200 
.15635 

.53981 
.55290 
.56564 

.02463 
.02607 
.02755 

.51513 
.52682 
.53809 

0.37 
0.38 
0.39 

.4477 
.4623 
.4770 

.69073 
.68386 
.67706 

0.3785 
0.3892 
0.4000 

1.0692 
1.0731 
1.0770 

.35399 
.36271 
.37136 

.16069 
.16503 
.16937 

.57807 
.59019 
.60202 

.02907 
.03063 
.03222 

.54899 
.55956 
.56980 

0.40 

1.4918 

.67032 

0.4108 

1.0811 

.37995 

0.17372 

1.61353 

0.03385 

1.67973 

0.41 
0.42 
0.43 

.5068 
.5220 
.5373 

.66365 
.65705 
.65051 

0.4216 
0.4325 
0.4434 

1.0852 
1.0895 
1.0939 

.38847 
.39693 
.40532 

.17806 
.18240 
.18675 

.62488 
.63594 
.64677 

.03552 
.03723 
.03897 

.58936 
.59871 
.60780 

0.44 
0.45 
0.46 

.5527 
.5683 
.5841 

.64404 
.  63763 
.63128 

0.4543 
0.4653 
0.4764 

1.0984 
1.1030 
1.1077 

.41364 
.42190 
.43008 

.19109 
.19543 
.19978 

.65738 
.66777 
.67797 

.04075 
.04256 
.04441 

.61663 
.62521 
.63355 

0.47 
0.48 
0.49 

.6000 
.6161 
.6323 

.62500 
.61878 
.61263 

0.4875 
0.4986 
0.5098 

1.1125 
1.1174 
1.1225 

.43820 
.44624 
.45422 

.20412 
.20846 
.21280 

.68797 
.69779 
.70744 

.04630 
.04822 
.05018 

.64167 
.64957 
.65726 

0.50 

1.6487 

.60653 

0.5211 

1.1276 

.46212 

0.21715 

1.71692 

0.05217 

1.66475 

0.51 
0.52 
0.53 

.6653 
.6820 
.6989 

.60050 
.59452 
.  58860 

0.5324 
0.5438 
0.5552 

1.1329 
1.1383 
1.1438 

.46995 
.47770 
.48538 

.22149 
.22583 
.23018 

.72624 
.73540 
.74442 

.05419 
.05625 
,05834 

.67205 
.67916 
.68603 

0.54 
0.55 
0.56 

.7160 
.7333 
.7507 

.58275 
.57695 
.57121 

0.5666 
0.5782 
0.5897 

1.1494 
1.1551 
1.1609 

.49299 
.50052 
.50798 

.23452 
.23886 
.24320 

.75330 
.76204 
.77065 

.  06046 
.06262 
.06481 

.69284 
.69942 
.70584 

0.57 
0.58 
0.59 

.7683 
.7860 
.8040 

.56553 
.55990 
.55433 

0.6014 
0.6131 
0.6248 

1.1669 
1.1730 
1.1792 

.51536 
.52267 
.52990 

.24755 
.25189 
.  25623 

.77914 
.78751 
.79576 

.  06703 
.06929 
.07157 

.71211 
.71822 
.72419 

0.60 

1.8221 

.54881 

0.6367 

1.1855 

.53705 

0.26058 

1.80390 

0.07389 

1.73001 

*  * 

Natural  Values 

Common  Logarithms 

e* 

e~* 

Sinks 

Cosh  a; 

Tanks 

e* 

Sinhz 

Cosh  re 

Tanha; 

0.60 

1.8221 

.54881 

0.6367 

1.1855 

.53705 

0.26058 

1.80390 

0.07389 

1.73001 

0.61 

1.8404 

.54335 

0.6485 

1,1919 

.54413 

.26492 

.81194 

.07624 

.  73570 

0  62 

1.8589 

.53794 

0.6605 

1.1984 

.55113 

.26926 

.81987 

«.  07861 

.74125 

0^63 

1.8776 

.53259 

0.6725 

1.2051 

.55805 

.27361 

.82770 

.08102 

.74667 

0.64 

1.8965 

.52729 

0.  6846 

1.2119 

.56490 

.27795 

.83543 

.08346 

.75197 

0  65 

1  9155 

.52205 

0.6967 

1.2188 

.57167 

.28229 

.84308 

.08593 

.75715 

Ol66 

1.9348 

.51685 

0.7090 

1.2258 

.57836 

.28663 

.85063 

.08843 

.76220 

0  67 

1.9542 

.51171 

0.7213 

1.2330 

.58498 

.29098 

.85809 

.09095 

.76714 

I  0*68 

1.9739 

.50662 

0.7336 

1.2402 

.59152 

.29532 

.86548 

.09351 

.77197 

0.69 

1.9937 

.50158 

0.7461 

1.2476 

.59798 

.29966 

.87278 

.09609 

.77669 

0.70 

2.0138 

.49659 

0.7586 

1.2552 

.60437 

0.30401 

1.88000 

0.09870 

1.78130 

0.71 

2.0340 

.49164 

0.7712 

1.2628 

.61068 

.30835 

.88715 

.10134 

.78581 

0  72 

2.0544 

.48675 

0.7838 

1.2706 

.61691 

.31269 

.89423 

.10401 

.79022 

0!73 

2.0751 

.48191 

0.7966 

1.2785 

.62307 

.31704 

.90123 

.10670 

.79453 

0.74 

2.0959 

.47711 

0.8094 

1.2865 

.62915 

.32138 

.90817 

.10942 

.79875 

0.75 

2.1170 

.47237 

0.8223 

1.2947 

.63515 

.32572 

.91504 

.11216 

.  80288 

0.76 

2.1383 

.46767 

0.8353 

1.3030 

.64108 

.33006 

.92185 

.11493 

.80691 

0.77 

2.  1598 

.46301 

0.8484 

1.3114. 

.64693 

.33441 

.92859 

,11773 

.81086 

0.78 

2.1815 

.45841 

0.8615 

1.3199 

.65271 

.33875 

.93527 

.12055 

.81472 

0,79 

2.2034 

.45384 

0.8748 

1.3286 

.65841 

.34309 

.94190 

.12340 

.81850 

0.80 

2.2255 

.44933 

0.8881 

1.3374 

.66404 

0.34744 

1.94846 

0.12627 

1.82219 

0.81 

2.2479 

.44486 

0.9015 

1.3464 

.66959 

.35178 

.95498 

.12917 

.82581 

0.82 

2.2705 

.44043 

0.9150 

1.3555 

.67507 

.35612 

.96144 

.13209 

82935 

0.83 

2.2933 

.43605 

0.9286 

1.3647 

.68048 

.36046 

.96784 

.13503 

.83281 

0.84 

2.3164 

.43171 

0.9423 

1.3740 

.68581 

.36481 

.97420 

.13800 

83620 

0.85 

2.3396 

.42741 

0.9561 

1.3835 

.69107 

.36915 

.98051 

.14099 

'83952 

0.86 

2.3632 

.42316 

0.9700 

1.3932 

.69626 

.37349 

.98677 

.14400 

,'84277 

0.87 

2.3869 

.41895 

0.9840 

1.4029 

.70137 

.37784 

.99299 

.14704 

.84595 

0.88 

2.4109 

.41478 

0.9981 

1.4128 

.70642 

.38218 

.99916 

.15009 

'  84906 

0.89 

2.4351 

.41066 

1.0122 

1.4229 

.71139 

.38652 

0.00528 

.15317 

185211 

0.90 

2.4596 

.40657 

1.0265 

1.4331 

,71630 

0.39087 

0.01137 

0.15627 

1.85509 

0.91 

2.4843 

.40252 

.0409 

1.4434 

.72113 

.39521 

.01741 

.15939 

.85801 

0.92 

2.5093 

.39852 

.0554 

1.4539 

.72590 

.39955 

.02341 

.16254 

'86088 

0.93 

2.5345 

.39455 

.0700 

1.4645 

.73059 

.40389 

.02937 

.16570 

186368 

0.94 
0.95 
0.96 

2.5600 
2.5857 
2.6117 

.39063 
.38674 
.38289 

.0847 
.0995 
.1144 

1.4753 
1.4862 
1.4973 

.73522 
.73978 
.74428 

.40824 
.41258 
.41692 

.03530 
.04119 
.04704 

.16888 
.17208 
.17531 

.86642 
.86910 
.87173 

0.97 
0.98 
0.99 

2.6379 
2.6645 
2.6912 

.37908 
.37531 
,37158 

.1294 
.1446 
.1598 

1.5085 
1.5199 
1.5314 

.74870 
.75307 
.75736 

.42127 
.42561 
.42995 

.05286 
.05864 
.06439 

.17855 

.18181 
.18509 

.87431 
.87683 
.87930 

1.00 

2.7183 

.36788 

.1752 

1.5431 

.76159 

0.43429 

0.07011 

0.18839 

1.88172 

.01 
.02 
.03 

2.7456 
2.7732 
2.8011 

.36422 
.36059 
.35701 

.1907 
.2063 
.2220 

1.5549 
1.5669 
1.5790 

.76576 
.76987 
.77391 

.43864 
.44298 
.44732 

.07580 
.08146 
.08708 

.19171 
.19504 
.19839 

.88409 
.88642 
.88869 

.04 
,  -05 
,06 

2.8292 
2.  8577 
2.8864 

.35345 
.34994 
.34646 

.2379 
.2539 
.2700 

1.5913 
1.6038 
1.6164 

.77789 
.78181 
.78566 

.45167 
.45601 
.46035 

.09268 
.09825 
.10379 

.20176 
.20515 
.20855 

.89092 
,89310 
.89524 

.07 
«  .08 

:  .09 

2.9154 
2.9447 
2.9743 

.34301 
.33960 
.33622 

.2862 
.3025 
.3190 

1.6292 
1.6421 
1.6552 

.78946 
.79320 
.79688 

.46470 
.46904 
.47338 

.10930 
.11479 
.12025 

.21197 
.21541 
.21886 

.89733 
.89938 
.90139 

1  .10 

3.0042 

.33287 

1.3356 

1.6685 

.80050 

0.47772 

0.12569 

0.22233 

1.90336 

»  .TI 
!  .12 
.13 

3.0344 
3.0649 
3.0957 

.32956 
,32628 
.32303 

.3524 
.3693 
.3863 

1.6820 
1.6956 
1.7093 

.80406 
.  80757 
.81102 

.48207 
.48641 
.49075 

.13111 
.13649 
.14186 

.22582 
.22931 
.23283 

.90529 
.90718 
.90903 

.14 
.15 
.16 

3.1268 
3.  1582 
3.1899 

.31982 
.31664 
.31349 

.4035 
.4208 
.4382 

1.7233 
1.7374 
1.7517 

.81441 
.81775 
.82104 

.49510 
.49944 
.50378 

.14720 
.15253 

.15783 

.23636 
.23990 
.24346 

,91085 
.91262 
.91436 

.17 
,18 
.19 

3.2220 
3.2544 
3.2871 

.31037 
.30728 
.30*22 

1.4558 
1.4735 
1.4914 

1.7662 
1.7808 
1.7957 

.82427 
.82745 
.83058 

,50812 
.51247 
.51681 

.16311 
.16836 
.17360 

.24703 
.25062 
.25422 

.91607 
.91774 
.91938 

1.20 

3.3201 

.30119 

1.5095 

1.8107 

.83365 

0.52115 

0,17882 

0.25784 

1.92099 

EXPONENTIAL  AND   HYPERBOLIC  TABLES 


1-29 


W 

Natural  Values 

Common  Logarithms 

e* 

e~x 

Sinha; 

Cosh  a; 

Tanha; 

ex 

Sinhre 

Cosh  a; 

Tanha; 

1.20 

3.3201 

.30119 

1.5095 

1.8107 

.83365 

0.52115 

0.17882 

0.25784 

1.92099 

.21 
.22 
.23 

3.3535 

3.3872 
3.4212 

.29820 
.29523 
.29229 

.5276 
.5460 
.5645 

.8258 
.8412 
.8568 

.83668 
.83965 
.84258 

.52550 
.52984 
.53418 

.18402 
.18920 
.19437 

.26146 
.26510 
.26876 

.92256 
.92410 
.92561 

.24 
1  .25 
.26 

3.4556 
3.4903 
3.5254 

.28938 
.28650 
.28365 

.5831 
.6019 
.6209 

.8725 
.8884 
.9045 

.84546 
.84828 
.85106 

.53853 
.54287 
.54721 

.19951 
.20464 
.20975 

.27242 
.27610 
.27979 

.92709 
.92854 
.92996 

.27 
.28 
.29 

3.5609 
3.5966 
3.6328 

.28083 
.27804 
.27527 

.6400 
.6593 
.6788 

.9208 
.9373 
.9540 

.85380 
.85648 
.85913 

.55155 
.55590 
.56024 

.21485 
.21993 
.22499 

.28349 
.28721 
.29093 

.93135 
.93272 
.93406 

.30 

3.6693 

.27253 

1.6984 

1.9709 

.86172 

0.56458 

0.23004 

0.29467 

1.93537 

.31 
.32 
.33 

3.7062 
3.7434 
3.7810 

.  26982 
.26714 
.26448 

.7182 
.7381 
.7583 

l'.9880 
2.0053 
2.0228 

.86428 
.86678 
.86925 

.56893 
.57327 
.57761 

.23507 
.24009 
.24509 

.29842 
.30217 
.30594 

.93665 
.93791 
.93914 

.34 
.35 
.36 

3.8190 
3.8574 
3.8962 

.26185 
.25924 
.25666 

.7786 
.7991 
.8198 

2.0404 
2.0583 
2.0764 

.87167 
.87405 
.87639 

.58195 
.58630 
.59064 

.25008 
.25505 
.26002 

.30972 
.31352 
.31732 

.94035 
.94154 
.94270 

.37 
.38 
.39 

3.9354 
3.9749 
4.0149 

.25411 
.25158 
.24908 

.8406 
.8617 
.8829 

2.0947 
2.1132 
2.1320 

.87869 
.88095 
.88317 

.59498 
.59933 
.60367 

.26496 
.26990 
.27482 

.32113 
.32495 
.32878 

.94384 
.94495 
.94604 

.40 

4.0552 

.24660 

1.9043 

2.1509 

.88535 

0.60801 

0.27974 

0.33262 

1.94712 

.41 
.42 
.43 

4.0960 
4.1371 
4  '787 

.24414 
.24171 
.23931 

.9259 
.9477 
.9697 

2.1700 
2.1894 
2.2090 

.88749 
.88960 
.89167 

.61236 
.61670 
.62104 

.28464 
.28952 
.29440 

.33647 
.34033 
.34420 

.94817 
.94919 
.95020 

.44 
.45 
.46 

4.2207 
4.2631- 
4.3060 

.23693 
.23457 
.23224 

1.9919 
2.0143 
2.0369 

2.2288 
2.2488 
2.2691 

.89370 
.89569 
.89765 

.62538 
.62973 
.63407 

.29926 
.30412 
.30896 

.34807 
.35196 
.35585 

.95119 
.95216 
.95311 

.47 

.48 
.49 

4.3492 
4.3929 
4.4371 

.22993 
.22764 
.  22537 

2.0597 
2.0827 
2.1059 

2.2896 
2.3103 
2.3312 

.89958 
.90147 
.90332 

.63841 
.64276 
.64710 

.31379 
.31862 
.32343 

.35976 
.36367 
.36759 

.95404 
.95495 
.95584 

.60 

4.4817 

.22313 

2.1293 

2.3524 

.90515 

0.65144 

0.32823 

0.37151 

1.95672 

.51 
.52 
.53 

4.5267 
4.5722 
4.6182 

.22091 
.21871 
.21654 

2.1529 
2.1768 
2.2008 

2.3738 
2.3955 
2.4174 

.90694 
.90870 
.91042 

.65578 
.66013 
.66447 

.33303 
.33781 
.34258 

.37545 
.37939 
.38334 

.95758 
.95842 
.95924 

.54 
.55 
.56 

4.6646 
4.7115 
4.7588 

.21438 
.21225 
.21014 

2.2251 
2.2496 
2.2743 

2.4395 
2.4619 
2.4845 

.91212 
.91379 
.91542 

.66881 
.67316 
.67750 

.34735 
.35211 
.35686 

.38730 
.39126 
.39524 

.96005 
.96084 
.96162 

.57 

.58 
.59 

4.8066 
4.8550 
4.9037 

.20805 
.20598 
.  20393 

2.2993 
2.3245 
2.3499 

2.5073 
2.5305 
2.5538 

.91703 
.91860 
.92015 

.68184 
.68619 
.  69053 

.36160 
.36633 
.37105 

.39921 
.40320 
.40719 

.96238 
.96313 
.96386 

1.60 

4.9530 

.20190 

2.3756 

2.5775 

.92167 

0.69487 

0.37577 

0.41119 

I.  96457 

.61 
.62 
.63 

5.0028 
5.0531 
5.1039 

.19989 
.19790 
.19593 

2.4015 
2.4276 
2.4540 

2.6013 
2.6255 
2.6499  • 

.92316 
.92462 
.92606 

.69921 
.70356 
.70790 

.38048 
.38518 
.38987 

.41520 
.41921 
.42323 

.96528 
.96597 
.96664 

.64 
.65 
.66 

5.1552 
5.2070 
5.2593 

.19398 
.19205 
.19014 

2.4806 
2.  5075 
2.5346 

2.6746 
2.6995 
2.7247 

.92747 
.92886 
.93022 

.71224 
.71659 
.72093 

.39456 
.39923 
.40391 

.42725 
.43129 
.43532 

.96730 
.96795 
,96858 

.67 
.68 
.69 

5.3122 
5.3656 
5.4195 

.18825 
.18637 
.18452 

2.5620 
2.5896 
2.6175 

2.7502 
2.7760 
2.8020 

.93155 
.93286 
.93415 

.72527 
.72961 
.73396 

.40857 
.41323 
.41788 

.43937 
.44341 
.44747 

.96921 

.96982 
.97042 

.70 

5.4739 

.18268 

2.6456 

2.8283 

.93541 

0.73830 

0.42253 

0.45153 

1.97100 

.71 
.72 
.73 

5.5290 
5.5845 
5.6407 

.18087 
.17907 
.17728 

2.6740 
2.7027 
2.7317 

2,8549 
2.8818 
2.9090 

.93665 
.93786 
.93906 

.74264 
.74699 
.75133 

.42717 
.43180 
.43643 

.45559 
.45966 
.46374 

.97158 
.97214 
.97269 

.74 
.75 
.76 

5.6973 
5.7546 
5.8124 

.17552 
.17377 
.17204 

2.7609 
2.7904 
2.8202 

2.9364 
2.9642 
2.9922 

.94023 
.94138 
.94250 

.75567 
.  76002 
.76436 

.44105 
.44567 
.45028 

.46782 
.47191 
.47600 

.97323 
.97376 
.97428 

.77 
.78 
.79 

5.8709 
5.9299 
5.9895 

.17033 
.  1  6864 
.16696 

2.8503 
2.  8806 
2.9112 

3.0206 
3.0492 
3.0782 

.94361 
.94470 
.94576 

.76870 
.77304 
.77739 

.45488 
.45948 
.  46408 

.48009 
.48419 
.48830 

.97479 
.97529 
.97578 

1.80 

6.0496 

.16530 

2.9422 

8.  1075 

.94681 

0.78173 

0.46867 

0.49241 

I.  97626 

1-30 


MATHEMATICS,   UNITS,   AND   SYMBOLS 


X 

Natural  Values 

Common  Logarithms 

& 

e~* 

Sinh  x 

Cosh  3 

Tanks 

e? 

Siuhz 

Coshx 

Tanha? 

1.80 

6.0496 

.16530 

2.9422 

3.1075 

.94681 

0.78173 

0.4686 

0.4924 

1.97626 

.81 

6  1104 

.16365 

2.9734 

3.1371 

.94783 

.7860 

.4732 

.4965 

.97673 

*82 

6"l719 

.16203 

3.0049 

3.1669 

.94884 

.79042 

.4778 

.5006 

.97719 

!83 

6.2339 

.16041 

3.0367 

3.1972 

.94983 

.79476 

.4824 

.5047 

.97764 

84 

6  2965 

.15882 

3.0689 

3.2277 

.95080 

.79910 

.4869 

.5088 

.97809 

85 

63598 

.15724 

3.1013 

3.2585 

.95175 

.  80344 

.49154 

.51302 

.97852 

.86 

6.4237 

.15567 

3.1340 

3.2897 

.95268 

.80779 

.49610 

.51716 

.97895 

.87 

6  4883 

.15412 

3.1671 

3.3212 

.95359 

.81213 

.50066 

.52130 

.97936 

88 

6*5535 

.15259 

3.2005 

3.3530 

.95449 

.81647 

.5052 

.52544 

.97977 

.89 

6,6194 

.15107 

3.2341 

3.3852 

.95537 

.82082 

.50976 

.52959 

.98017 

.90 

6.6859 

.14957 

3.2682 

3.4177 

.95624 

0.82516 

0.51430 

0.53374 

1.98057 

.91 

6.7531 

.14808 

3.3025 

3.4506 

.95709 

.82950 

.51884 

.53789 

.98095 

92 

6.8210 

.14661 

3.3372 

3.4838 

.95792 

.83385 

.52338 

.54205 

.98133 

.93 

6.8895 

.14515 

3.3722 

3.5173 

.95873 

.83819 

.52791 

.54621 

.98170 

94 

6.9588 

.14370 

3.4075 

3.5512 

.95953 

.  84253 

.53244 

.55038 

.98206 

;   .95 

7.0287 

.14227 

3.4432 

3.5855 

.96032 

.84687 

.53696 

.55455 

.98242 

.96 

7.0993 

.14086 

3.4792 

3.6201 

.96109 

.85122 

.54148 

.55872 

.98272 

.97 

7.1707 

.  13946 

3.5156 

3.6551 

.96185 

.85556 

.54600 

.56290 

.98311 

.98 

7.2427 

.13807 

3.5523 

3.6904 

.96259 

.85990 

.55051 

.56707 

.98344 

.99 

7.3155 

.13670 

3.5894 

3.7261 

.96331 

.  86425 

.55502 

.57126 

.98377 

2.00 

7.3891 

.13534 

3.6269 

3.7622 

.96403 

0.86859 

0.55953 

0.57544 

1.98409 

'  2.01 

7.4633 

.13399 

3.6647 

3.7987 

.96473 

.87293 

.56403 

.  .57963 

.98440 

2.02 

7.5383 

.  13266 

3.7028 

3.8355 

.96541 

.87727 

.56853 

.58382 

.98471 

2.03 

7.6141 

.13134 

3.7414 

3.8727 

.96609 

.88162 

.57303 

.58802 

.96502 

2.04 

7.6906 

.13003 

3.7803 

3.9103 

.96675 

.88596 

.57753 

.59221 

.98531 

2.05 

7.7679 

.12873 

3.8196 

3.9483 

.96740 

.89030 

.58202 

.59641 

.98560 

2.06 

7.8460 

.12745 

3.8593 

3.9867 

.96803 

.89465 

.  58650 

.60061 

.98589 

2.07 

7,9248 

.12619 

3.8993 

4.0255 

.96865 

.89899 

.59099 

.60482 

.98617 

2.08 

8.0045 

.12493 

3.9398 

4.0647 

.96926 

.  90333 

.59547 

.  60903 

.98644 

2.09 

8.0849 

.12369 

3.9806 

4.1043 

.96986 

.90768 

.  59995 

.61324 

.98671 

2.10 

8.1662 

.12246 

4.0219 

4.1443 

.97045 

0.91202 

0.60443 

0.61745 

1.98697 

2.11 

8.2482 

.12124 

4.0635 

4.1847 

.97103 

.91636 

.  60890 

.62167 

.98723 

2.12 

8.3311 

.12003 

4.1056 

4.2256 

.97159 

.92070 

.61337 

.62589 

.98748 

2.13 

8.4149 

.11884 

4.  1480 

4.2669 

.97215 

.92505 

.61784 

.63011 

.98773 

2.14 

8.4994 

.11765 

4.1909 

4.3085 

.97269 

.92939 

.62231 

.  63433 

.  98798 

2.15 

8.5849 

.11648 

4.2342 

4.3507 

.97323 

.93373 

.  62677 

.63856 

.98821 

2.16 

8.6711 

.11533 

4.2779 

4.3932 

.97375 

.93808 

.63123 

.64278 

.98845 

2.T7 

8.7583 

.11418 

4.3221 

4.4362 

.97426 

.94242 

.63569 

.64701 

.98868 

2.18 

8.8463 

.11304 

4.3666 

4.4797 

.97477 

.94676 

.64015 

.65125 

.98890 

2.19 

8.9352 

.11192 

4.4116 

4.5236 

.97526 

.95110 

.64460 

.  65548 

.98912 

2.20 

9.0250 

.11080 

4.4571 

4.5679 

.97574 

0.95545 

0.64905 

0.65972 

1.98934 

2.21 

9.1157 

.10970 

4.5030 

4.6127 

.97622 

.95979 

.65350 

.66396 

.  98955 

2.22 

9.2073 

.10861 

4.5494 

4.6580 

,97668 

.96413 

.65795 

.66820 

98975 

2.23 

9.2999 

.10753 

4.5962 

4.7037 

,97714 

.96848 

.66240 

.  67244 

.98996 

2.24 

9.3933 

.10646 

4.6434 

4.7499 

.97759 

.97282 

.  66684 

.  67668 

99016 

2.25 
2.26 

9.4877 
9.5831 

.10540 
.10435 

4.6912 
4.7394 

4.7966 
4.8437 

.97803 
.97846 

.97716 
.98151 

.67128 
.67572 

.  68093 
.68518 

.99035 
.99054 

2.27 
2.28 
2.29 

9.6794 
9.7767 
9.8749 

.  10331 
.10228 
.10127 

4.7880 
4.8372 
4.8868 

4.8914 
4.9395 
4.9881 

.  97888 
.  97929 
.97970 

.98585 
.99019 
.99453 

.68016 
.  68459 
.68903 

.68943 
.  69368 
.69794 

.99073 
.99091 
.99109 

2.30 

9.9742 

.10026 

4.9370 

5.0372 

.98010 

0.99888 

0.69346 

0.70219 

.99127 

2.3t 

2.32 
2.33 

10.074 
10.176 
10.278 

.09926 
.09827 
.09730 

4.9876 
5.0387 
5.0903 

5.0868 
5.  1370 
5.1876 

.98049 
.98087 
.98124 

.  00322 
.00756 
.01191 

.69789 
.  70232 
.70675 

,70645 
.71071 
.71497 

.99144 
.99161 
.99178 

2.34 
2.35 
2.36 

10.381 
10.486 
10.591 

.09633 
.09537 
.09442 

5.1425 
5.1951 
5.2483 

5.2388 
5.2905 
5.3427 

.98161 
.98197 
.98233 

.01625 
.02059 
.02493 

.71117 
.71559 
.72002 

.71923 
.72349 
.72776 

.99194 
.99210 
.99226 

2.37 
2.38 
2.39 

10.697 
10.805 
10.913 

.09348 
.09255 
.09163 

5.3020 
5.3562 
5.4109 

5.3954 
5.4487 
5.5026 

.98267 
.98301 
.98335 

.02928 
.03362 
.03796 

.72444 
.  72885 
.  73327 

.73203 
.  73630 
.74056 

.99241 
.99256 
.99271 

1.40 

U.Ott 

.09072 

5.4662 

9.5569 

.98367 

.04231 

.73769 

.74484 

.99285 

EXPONENTIAL  AND  HYPERBOLIC  TABLES 


1-31 


X 

Natural  Values 

Common  Logarithms 

<P 

e~x 

Sinhz 

Cosh  a; 

Tanhx 

ex 

Sinha; 

Cosh  a; 

Tanha; 

2.40 

11.023 

.09072 

5.4662 

5.5569 

.98367 

1.04231 

0.73769 

0.74484 

1.99285 

2.41 
2.42 
2.43 

11.134 
11.246 
11.359 

.08982 
.08892 
.08804 

5.5221 
5.5785 
5.6354 

5.6119 
5.6674 
5.7235 

.98400 
.98431 
.98462 

.04665 
.05099 
.05534 

.74210 
.74652 
.75093 

.74911 
.75338 
.75766 

.99299 
.99313 
.99327 

2.44 
2.45 
2.46 

11.473 
11.588 
11.705 

.08716 
.08629 
.08543 

5.6929 
5.7510 
5.8097 

5.7801 
5.8373 
5.8951 

.98492 
.98522 
.98551 

.05968 
.06402 
.06836 

.75534 
.75975 
.76415 

.76194 
.76621 
.77049 

.99340 
.99353 
.99366 

2.47 
2.48 
2.49 

11.822 
11.941 
12.061 

.08458 
.08374 
.08291 

5.8689 
5.9288 
5.9892 

5.9535 
6.0125 
6.0721 

.98579 
.98607 
.98635 

.07271 
.07705 
.08139 

.76856 
.77296 
.  77737 

.77477 
.77906 
.78334 

.99379 
.99391 
.99403 

2.50 

12.182 

.08208 

6.0502 

6.1323 

.98661 

1.08574 

0.78177 

0.78762 

1.99415 

2.51 
2.52 
2.53 

12.305 
12.429 
12.554 

.08127 
.08046 
.07966 

6.1118 
6.1741 
6.2369 

6.1931 
6.2545 
6.3166 

.98688 
.98714 
.98739 

.09008 
.09442 
.09877 

.78617 
.  79057 
.79497 

.79191 
.79619 
.80048 

.99426 
.99438 
.99449 

2.54 
2.55 
2.56 

12.680 
12.807 
12.936 

.07887 
.07808 
.07730 

6.3004 
6.3645 
6.4293 

6.3793 
6.  4426 
6.5066 

.98764 
.98788 
.98812 

.10311 
.10745 
.11179 

.79937 
.80377 
.80816 

.80477 
.80906 
.81335 

.99460 
.99470 
.99481 

2.57 
2.58 
2.59 

13.066 
13.197 
13.330 

.07654 
.07577 
.07502 

6.4946 
6.5607 
6.6274 

6.5712 
6.  6365 
6.7024 

.98835 
.98858 
.98881 

.11614 
.12048 
.12482 

.81256 
.81695 
.82134 

.81764 
.82194 
.82623 

.99491 
.99501 
.99511 

2.60 

13.464 

.07427 

6.6947 

6.7690 

.98903 

1.12917 

0.82573 

0.83052 

1.99521 

2.61 
2.62 
2.63 

13.599 
13.736 
13.874 

.07353 
.07280 
.07208 

6.7628 
6.8315 
6.9008 

6.8363 
6.9043 
6.9729 

.98924 
.98946 
.98966 

.13351 
.13785 
.14219 

.83012 
.83451 
.83890 

.83482 
.83912 
.84341 

.99530 
.99540 
.99549 

2.64 
2.65 
2.66 

14.013 
14.154 
14.296 

.07136 
.07065 
.06995 

6.9709 
7.0417 
7.1132 

7.0423 
7.  1123 
7.1831 

.98987 
.99007 
.99026 

.14654 
.15088 
.15522 

.  84329 
,  84768 
.  85206 

.84771 
.85201 
.85631 

.99558 
.99566 
.99575 

2.67 
2.68 
2.69 

14.440 
14.585 
14.732 

.06925 
.06856 
.06788 

7.1854 
7.2583 
7.3319 

7.2546 
7.3268 
7.3998 

.99045 
.99064 
.99083 

.15957 
.16391 
.16825 

.  85645 
.  86083 
.  86522 

.86061 
.86492 
.86922 

.99583 
.99592 
.99600 

2.70 

14.880 

.06721 

7.4063 

7.4735 

.99101 

1.17260 

0.86960 

0.87352 

1.99608 

2.71 
2.72 
2.73 

15.029 
15.180 
15.333 

.06654 
.06587 
.06522 

7.4814 
7.5572 
7.6338 

7.5479 
7.6231 
7.6991 

.99118 
.99136 
.99153 

.17694 
.18128 
.18562 

.  87398 
.87836 
.88274 

.87783 
.88213 
.88644 

.99615 
.99623 
.99631 

2.74 
2.75 
2.76 

15.487 
15.643 
15.800 

.06457 
.06393 
.  06329 

7.7112 
7.7894 
7.8683 

7.7758 
7.8533 
7.9316 

.99170 
.99186 
.99202 

.18997 
.19431 
.19865 

.88712 
.89150 
.  89588 

.89074 
.89505 
.89936 

.99638 
.99645 
.99652 

2.77 
2.78 
2.79 

15.959 
16.119 
16.281 

.06266 
.06204 
.06142 

7.9480 
8.0285 
8.1098 

8.0106 
8.0905 
8.1712 

.99218 
.99233 
.99248 

.20300 
.20734 
.21168 

.  90026 
.  90463 
.90901 

.90367 
.90798 
.91229 

.99659 
.99666 
.99672 

2.80 

16.445 

.06081 

8.1919 

8.2527 

.99263 

1.21602 

0.91339 

0.91660 

1.99679 

2.81 
2.82 
2.83 

16.610 
16.777 
16.945 

.06020 
.05961 
.05901 

8.2749 
8.3586 
8.4432 

8.3351 
8.4182 
8.5022 

.99278 
.99292 
.99306 

.22037 
.22471 
.22905 

.91776 
.92213 
.92651 

.92091 
.92522 
.92953 

.99685 
.99691 
.99698 

2.84 
2.85 
2.86 

17.116 
17.288 
17.462 

.05843 
.05784 
.05727 

8.5287 
8.6150 
8.7021 

8.5871 
8.6728 
8.7594 

.99320 
.99333 
.99346 

.23340 
.23774 
.24208 

.93088 
.  93525 
.93963 

.93385 
.93816 
.94247 

.99704 
.99709 
.99715 

2.87 
2.88 
2.89 

17.637 
17.814 
17.993 

.  05670 
.05613 
.05558 

8.7902 
8.8791 
8.9689 

8.8469 
8.9352 
9.0244 

.  99359 
.99372 
.99384 

.24643 
.25077 
.25511 

.  94400 
.94837 
.95274 

.94679 
.95110 
.95542 

.99721 
.99726 
.99732 

2.90 

18.174 

.05502 

9.0596 

9.1146 

.99396 

1.25945 

0.95711 

0.95974 

T.  99737 

2.91 
2.92 
2.93 

18.357 
18.541 
18.728 

.05448 
.05393 
.05340 

9.1512 
9.2437 
9.3371 

9.2056 
9.2976 
9.3905 

.99408 
.99420 
.99431 

.26380 
.26814 
.27248 

.96148 
.96584 
.97021 

.96405 
.96837 
.  97269 

.99742 
.99747 
.99752 

2.94 
2.95 
2.96 

18.916 
19.106 
19.298 

.05287 
.05234 
.05182 

9.4315 
9.5268 
9.6231 

9.4844 
9.5791 
9.6749 

.99443 
.99454 
.99464 

.27683 
.28117 
.28551 

.97458 
.97895 
.98331 

.97701 
.98133 
.98565 

.99757 
.99762 
.99767 

2.97 
2.98 
2.99 

19.492 
19.688 
19.886 

.05130 
.05079 
.05029 

9.7203 
9.8185 
9.9177 

9.7716 
9.8693 
9.9680 

.99475 
.99485 
.99496 

.28985 
.29420 
.  29854 

.98768 
.99205 
.99641 

.98997 
.99429 
,99861 

.99771 
.99776 
.99780 

3.00 

20.086 

.04979 

10.018 

10.068 

.99505 

1.30283 

1.00078 

1.00293 

1.99785 

1-32 


MATHEMATICS,  UNITS,  AND   SYMBOLS 


EXPONENTIAL  AND   HYPERBOLIC  TABLES 


1-33 


X 

Natural  Values 

Common  Logarithms 

& 

e~* 

Sinha; 

Cosh  a; 

Tanhrc 

€* 

Sinha 

Cosh  a; 

Tanha: 

3.60 

36.598 

.02732 

18.285 

18.313 

.99851 

1.56346 

1.26211 

1.26275 

1.99935 

3.61 
3.62 
3.63 

36.966 
37.338 
37.713 

.02705 
.02678 
.02652 

18.470 
18.655 
18.843 

18.497 
18.682 
18.870 

.99854 
.99857 
.99859 

.56780 
.57215 
.57649 

.26646 
.27080 
.27515 

.26709 
.27143 
.27576 

.99936 
.99938 
.99939 

3.64 
3.65 
3.66 

38.092 

38.475 
38.861 

.02625 
.02599 
.02573 

19.033 
19.224 
19.418 

19.059 
19.250 
19.444 

.99862 
.99865 
.99868 

,58083 
.58517 
.58952 

.27950 
.28385 
.28820 

.28010 
.28444 
.28878 

.99940 
.99941 
.99942 

3.67 
3.68 
3.69 

39.252 
39.646 
40.045 

.02548 
.02522 
.02497 

19.613 
19.811 
20.010 

19.639 
19.836 
20.035 

.99870 
.99873 
.99875 

.59386 
.59820 
.60255 

.29255 
.29690 
.30125 

.29311 
.29745 
.30179 

.99944 
.99945 
.99946 

3.70 

40.447 

.02472 

20.211 

20.236 

.99878 

1.60689 

1.30559 

1.30612 

1.99947 

3.71 
3.72 
3.73 

40.854 
41.264 
41.679 

.02448 
.02423 
.02399 

20.415 
20.620 
20.828 

20.439 
20.644 
20.852 

.  99'880 
.99883 
.99885 

.61123 
.61558 
.61992 

.30994 
.31429 
.31864 

.31046 
.31480 
.31914 

.99948 
.99949 
.99950 

3.74 
3.75 
3.76 

42.098 
42.521 
42.948 

.02375 
.02352 
.02328 

21.037 
21.249 
21.463 

21.061 
21.272 
21.486 

.99887 
.99889 
.99892 

.62426 
.62860 
.63295 

.32299 
.32733 
.33168 

.32348 
.32781 
,33215 

.99951 
.99952 
.99953 

3.77 

3.78 
3.79 

43.380 
43.816 
44.256 

.02305 
.02282 
.02260 

21.679 
21.897 
22.117 

21.702 
21.919 
22.140 

.99894 
.99896 
.99898 

.63729 
.64163 
.64598 

.33603 
.34038 
.34472 

.33649 
.34083 
.34517 

.99954 
.99955 
.99956 

3.80 

44.701 

.02237 

22.339 

22.362 

.99900 

1.65032 

1.34907 

1.34951 

1.99957 

3.81 
3.82 
3.83 

45.150 
45.604 
46.063 

.02215 
.02193 
.02171 

22.564 
22.791 
23.020 

22.586 
22.813 
23.042 

.99902 
.99904 
.99906 

.65466 
.65900 
.66335 

.35342 
.35777 
.36211 

\35384 

.35818 
.36252 

.99957 
.99958 
.99959 

3.84 
3.85 
3,86 

46.525 
46.993 
47.465 

.02149 
.02128 
.02107 

23.252 
23.486 
23.722 

23.274 
23.507 
23.743 

.99908 
.99909 
.99911 

.66769 
.67203 
.67638 

.36646 
.37081 
.37515 

.36686 
.37120 
.37554 

.99960 
.99961 
.99961 

3.87 
3.88 
3.89 

47.942 
48.424 
48.911 

.02086 
.02065 
.02045 

23.961 
24.202 
24.445 

23.982 
24.222 
24.466 

.99913 
.99915 
.99916 

.68072 
.68506 
.68941 

.37950 
.38385 
.38819 

.37988 
.38422 
.38856 

.99962 
.99963 
.99964 

3.90 

49.402 

.02024 

24.691 

24.711 

.99918 

1.69375 

1.39254 

1.39290 

1.99964 

3.91 
3.92 
3.93 

49.899 
50.400 
50.907 

.02004 
.01984 
.01964 

24,939 
25.190 
25.444 

24.960 
25.210 
25.463 

.99920 
.99921 
.99923 

.69809 
.70243 
.70678 

.39689 
.40123 
.40558 

.39724 
.40158 
.40591 

.99965 
.99966 
.99966 

3.94 
3.95 
3.96 

51.419 
51.935 
52.457 

.01945 
.01925 
.01906 

25.700 
25.958 
26.219 

25.719 
25.977 
26.238 

.99924 
.99926 
.99927 

.71112 
.71546 
.71981 

.40993 

.41427 
.41862 

.41025 
.41459 
.41893 

.99967 
.99968 
.99968 

3.97 
3.98 
3.99 

52.985 
53.517 
54.055 

.01887 
.01869 
.01850 

26.483 
26.749 
27.018 

26.502 
26.768 
27.037 

.99929 
.99930 
.99932 

.72415 
.72849 
.73284 

.42296 
.42731 
.43166 

.42327 
.42761 
.43195 

.99969 
.99970 
.99970 

4.00 

54.598 

.01832 

27.290 

27.308 

.99933 

1.73718 

1.43600 

1.43629 

1.99971 

4.01 
4.02 
4.03 

55.147 
55.701 
56.261 

.01813 
.01795 
.01777 

27.564 
27.842 
28.122 

27.583 
27.860 
28.139 

.99934 
.99936 
.99937 

.74152 
.74586 
.75021 

.44035 
.44469 
.44904 

.44063 
.44497 
.44931 

.99971 
.99972 
.99973 

4.04 
4.05 
4.06 

56.826 
57.397 
57.974 

.01760 
.01742 
.01725 

28.404 
28.690 
28.979 

28.422 
28.707 
28.996 

.99938 
.99939 
.99941 

.75455 

.75889 
.76324 

.45339 
.45773 
.46208 

.45365 
.45799 
.46233 

.  99973 
.99974 
.99974 

4.07 
4.08 
4.09 

58.557 
59.145 
59.740 

.01708 
.01691 
.01674 

29.270 
29.564 
29.862 

29.287 
29.581 
29.878 

.99942 
.99943 
.99944 

.76758 
.77192 
.77626 

.46642 
.47077 
.47511 

.46668 
.47102 
.47536 

.99975 
.99975 
.99976 

4.10 

60.340 

.01657 

30.162 

30.178 

.99945 

1.78061 

1.47946 

1,47970 

1.99976 

4.11 
4.12 
4.13 

60.947 
61.559 
62.178 

.01641 
.01624 
.01608 

30.465 
30.772 
31.081 

30.482 
30.788 
31.097 

.99946 
.99947 
.99948 

.78495 
.78929 
.79364 

.48380 
.48815 
.49249 

.48404 
.48838 
.49272 

.99977 
.99977 
.99978 

4.14 
4.15 
4.16 

62.803 
63.434 
64.072 

.01592 
.01576 
.01561 

31-.393 
31.709 
32.028 

31.409 
31.725 
32.044 

.99949 
.99950 
.99951 

.79798 
.80232 
.80667 

.49684 
.50118 
.50553 

.49706 
.50140 
.50574 

.99978 
.99978 
.99979 

4.17 
4.18 
4.19 

64.715 
65.366 
66.023 

.01545 
.01530 
.01515 

32.350 
32.675 
33.004 

32.365 
32.691 
33.019 

.99952 
.99953 
.99954 

.81101 
.81535 
.81969 

.50987 
.51422 
.51856 

.51008 
.51442 
.51876 

.99979 
.99980 
.99980 

4.20 

66.686 

.01500 

33.336 

33.351 

.99955 

1.82404 

1.52291 

1.52310 

1.99980 

1-34 


MATHEMATICS,  UNITS,  AND  SYMBOLS 


X 

Natural  Values 

Common  Logarithms 

<F 

e-* 

Sinks 

Cosh  s 

Tanhre 

e° 

Sinha; 

Cosh  a; 

Tanha: 

4.20 

66.686 

.01500 

33.336 

33.35 

.99955 

1.82404 

1.52291 

1.52310 

1.99980 

4.21 

67.357 

.01485 

33.67 

33.68 

.99956 

.82838 

.52725 

.52745 

.99981 

4.22 

68.033 

.01470 

34.009 

34.02 

.99957 

.83272 

.53160 

.53179 

.99981 

4.23 

68.717 

.01455 

34.351 

34.36 

.99958 

.83707 

.53594 

.53613 

.99982 

4.24 

69.408 

.01441 

34.697 

34.71 

.99958 

.84141 

.54029 

.54047 

.99982 

4.25 

70.105 

.01426 

35.046 

35.060 

.99959 

.84575 

.54463 

.54481 

.99982 

4.26 

70.810 

.01412 

35.398 

35.412 

.99960 

.85009 

.54898 

.54915 

.99983 

4.27 

71.522 

.01398 

35.754 

35.768 

.99961 

.  85444 

.55332 

.55349 

.99983 

4.28 

72.240 

.01384 

36.113 

36.127 

.99962 

.85878 

.55767 

.55783 

.99983 

4.29 

72.966 

.01370 

36.476 

36.490 

.99962 

.86312 

.56201 

.56217 

.99984 

4.30 

73.700 

.01357 

36.843 

36.857 

.99963 

1.86747 

1.66636 

1.56652 

1.99984 

4.31 

74.440 

.01343 

37.214 

37.227 

.99964 

.87181 

.57070 

.57086 

.99984 

4.32 

75.189 

.01330 

37.588 

37.60 

.99965 

.87615 

.57505 

.57520 

.99985 

4.33 

75.944 

.01317 

37.966 

37.979 

.99965 

.  88050 

.57939 

.57954 

.99985 

4.34 

76.708 

.01304 

38.347 

38.360 

.99966 

.  88484 

.58373 

.58388 

.99985 

4.35 

77.478 

.01291 

38.733 

38.746 

.99967 

.88918 

.58808 

.58822 

99986 

4.36 

78.257 

.01278 

39.122 

39.135 

.99967 

.89352 

.59242 

.59256 

.99986 

4.37 

79.044 

.01265 

39.515 

39.528 

.99968 

.  89787 

.59677 

.59691 

.99986 

4.38 

79.838 

.01253 

39.913 

39.925 

.99969 

.90221 

.60111 

.60125 

99986 

4.39 

80.640 

.01240 

40.314 

40.326 

.99969 

.90655 

.60546 

.60559 

.99987 

4.40 

81.451 

.01228 

40.719 

40.732 

.99970 

1.91090 

1.60980 

1.60993 

1.99987 

4.47 
4.42 
4.43 

82.269 
83.096 
83.931 

.01216 
.01203 
.01191 

41.129 
41.542 
41.960 

41.141 
41.554 
41.972 

.99970 
.99971 
.99972 

.91524 
.91958 
.92392 

.61414 
.61849 
.62283 

.61427 
.61861 
.  62296 

.99987 
.99987 
.99988 

4.44 
4.45 
4.46 

84.775 
85.627 
86.488 

.01180 
.01168 
.01156 

42.382 
42.808 
43.238 

42.393 
42.819 
43.250 

.99972 
.99973 
.99973 

'.92827 
.93261 
.  93695 

.62718 
.63152 
.63587 

.  62730 
.  63  1  64 
.  63598 

.99988 
.99988 
.99988 

4.47 
4.48 
4.49 

87.357 
88.235 
89.121 

.01145 
.01133 
.01122 

43.673 
44.112 
44.555 

43.684 
44.123 
44.566 

.99974 
.99974 
.99975 

.94130 
.94564 
.94998 

.64021 
.64455 
.  64890 

.  64032 
.64467 
.64901 

.99989 
.99989 
.99989 

4.50 

90.017 

.01111 

45.003 

45.014 

.99975   1.95433 

1.65324 

1.65335 

1.99989 

4.51 
4.52 
4.53 

90.  922 
91.836 
92.759 

.01100 
.01089 
.01078 

45.455 
45.912 
46.374 

45.466 
45.923 
46.385 

.99976 
.99976 
.99977 

.  95867 
.96301 
.  96735 

.65759 
.66193 
.  66627 

.65769 
.  66203 
.  66637 

.99989 
.99990 
.99990 

4.54 
4.55 
4.56 

93.691 
94.632 
95.583 

.01067 
.01057 
.01046 

46.840 
47.311 
47.787 

46.851 
47.321 
47.797 

.99977 
.99978 
.99978 

.97170 
.  97604 
.98038 

.  67062 
.67496 
.67931 

.67072 
.  67506 
.67940 

.  99990 
.99990 
.99990 

4.57 
4.58 
4.59 

96.544 
97.514 
98.494 

.01036 
.01025 
.01015 

48.267 
48.752 
49.242 

48.277 
48.762 
49.252 

.99979 
.99979 
.99979 

.98473 
.98907 
.99341 

.68365 
.68799 
.69234 

.  68374 
.68808 
.69243 

.99991 
.99991 
.99991 

4.60 

99.484 

.01005 

49.737 

49.747 

.99980 

1.99775 

1.69668 

1.69677 

1.99991 

4.61 
4.62 
4.63 

100.48 
101,49 
02.51 

.00995 
.00985 
.00975 

50.237 
50.742 
51.252 

50.247 
50.752 
51.262 

.99980   2.00210 
.99981    .00644 
.99981    .01078 

.70102 
.70537 
.70971 

.70111 
.70545 
.70979 

.99991 
.99992 
.99992 

4.64 
4.65 
4.66 

03.54 
04.58 
05.64 

.  00966 
.00956 
.00947 

51.767 
52.288 
52.813 

51.777 
52.297 
52.823 

.99981    .01513 
.99982    .01947 
.99982    .02381 

.71406 
.71840 
.72274 

.71414 
.71848 
.72282 

.99992 
.99992 
.99992 

4.67 
4.68 
4.69 

06.70 
07.77 
08.85 

.00937 
.00928 
.00919 

53.344 
53.880 
54.422 

53.354 
53.890 
54.431 

.99982    .02816 
.99983    .03250 
.99983    .03684 

.72709 
.73143 
.73577 

.72716 
.73151 
.73585 

.99992 
.99993 
.99993 

4.70 

109.95 

.00910 

54.969 

54.978 

.99983   2.04118 

1.74012 

1.74019 

1.99993 

4.71 
4.72 
4.73 

11.05 
12.17 
13.30 

.00900 
.00892 
.00883 

55.522 

56.080 
56.643 

55.531 
56.089 
56.  652 

.99984    .04553 
.99984    .04987 
.99984    .05421 

.74446 
.74881 
.75315 

.  74453 
.  74887 
.75322 

.99993 
.99993 
.  99993 

4.74 
4.75 
4.76 

14.43 
15.58 
16.75 

.00874 
.00865 
.00857 

57.213 
57.788 
58.369 

57.222 
57.796 
58.377 

.99985    .05856- 
.99985    .06290 
.99985    .06724 

.75749 
.76184 
.76618 

.75756 
.76190 
.  76624 

.99993 
.99993 
.99994 

4.77 
4.78 
4.79 

17.92 

19.10 
20.30 

.00848 
.00840 
.00831 

58.955 
59.548 
60.147 

58.964 
59.556 
60.155 

99986    .07158 
99986    .07593 
99986    .08027 

.77052 
.77487 
.77921 

.77059 
.77493 
.77927 

.99994 
.99994 
.99994 

4.80 

121.51 

.00823 

60.751 

60.759 

99986   2.08461 

1.78355 

1.78361  i 

L.  99994 

EXPONENTIAL  AND  HYPERBOLIC  TABLES 


1-35 


X 

Natural  Values] 

Common  Logarithms 

e* 

e-s 

Sinho; 

Cosh  a; 

Tauhs 

«* 

Sinhs 

Cosh  a; 

Tanh  x 

4.80 

121.51 

.00823 

60.751 

60.760 

.99986 

2.08461 

1.78355 

1.78361 

1.99994 

4.81 
4.82 
4.83 

122.73 
123.97 
125.21 

.00815 
.00807 
.00799 

61.362 
61.979 
62.601 

61.370 
61.987 
62.609 

.99987 
.99987 
.99987 

.08896 
.09330 
.09764 

.78790 
.79224 
.79658 

.78796 
.79230 
.79664 

.99994 
.99994 
.99994 

4.84 
4.85 
4.86 

126.47 
127.74 
129.02 

.00791 
.00783 
.00775 

63.231 
63.866 
64.508 

63.239 
63.874 
64.516 

.99987 
.99988 
.99988 

.10199 
.10633 
.11067 

.80093 
.80527 
.80962 

.80098 
.80532 
.80967 

.99995 
.99995 
.99995 

4.87 
4.88 
4.89 

130.32 
131.63 
132.95 

.00767 
.00760 
,00752 

65.157 
65.812 
66.473 

65.164 
65.819 
66.481 

.99988 
.99988 
.99989 

.11501 
.11936 
.12370 

.81396 
.81830 
.82265 

.81401 
.81835 
.82269 

.99995 
.99995 
.99995 

4.90 

134.29 

.00745 

67.141 

67.149 

.99989 

2.12804 

1.82699 

1.82704 

1.99995 

4.91 
4.92 
4.93 

135.64 
137.00 
138.38 

.00737 
.00730 
.00723 

67.816 
68.498 
69.186 

67.823 
68.505 
69.193 

.99989 
.99989 
.99990 

.13239 
.13673 
.14107 

.83133 
.83568 
.84002 

.83138 
.83572 
.84006 

.99995 
.99995 
.99995 

4.94 
4.95 
4.96 

139.77 
141.17 
142.59 

.00715 
.00708 
.00701 

69.882 
70.584 
71.293 

69.889 
70.591 
71.300 

.99990 
.99990 
.99990 

.14541 
.14976 
.15410 

.84436 
.84871 
.85305 

.84441 
.84875 
.85309 

.99996 
.99996 
,99996 

4.97 
4.98 
4.99 

144.03 
145.47 
146.94 

.00694 
.00687 
.00681 

72.010 
72.734 
73.465 

72.017 
72.741 
73.472 

.99990 
.99991 
.99991 

.15844 
.16279 
.16713 

.85739 
.86174 
.86608 

.85743 
.86178 
.86612 

.99996 
.99996 
.99996 

6.00 

148.41 

.00674 

74.203 

74.210 

.99991 

2.17147 

1.87042 

1.87046 

1.99996 

5.01 
5.02 
5.03 

149.90 
151.41 
152.93 

.00667 
.00660 
.00654 

74.949 
75.702 
76.463 

74.956 
75.710 
76.470 

.99991 
.99991 
.99991 

.17582 
.18016 
.18450 

.87477 
.87911 
.88345 

.87480 
.87915 
.88349 

.99996 
.99996 
.99996 

5.04 
5.05 
5.06 

154.47 
156.02 
157.59 

.00647 
.00641 
.00635 

77.232 
78.008 
78.792 

77.238 
78.014 
78.798 

.99992 
.99992 
.99992 

.18884 
.19319 
.19753 

.88780 
.89214 
.89648 

.88783 
.89217 
.89652 

.99996 
.99996 
.99997 

5.07 
5.08 
5.09 

159.17 
160.77 
162.39 

.00628 
.00622 
.00616 

79.584 
80.384 
81.192 

79.590 
80.390 
81.198 

.99992 
.99992 
.99992 

.20187 
.20622 
.21056 

.90083 
.90517 
.90951 

.90086 
.90520 
.90955 

.99997 
.99997 
.99997 

6.10 

164.02 

.00610 

82.008 

82.014 

.99993 

2.21490 

1.91386 

1.91389 

1.99997 

5.11 
5.12 
5.13 

165.67 
167.34 
169.02 

.00604 
.00598 
.00592 

82.832 
83.665 
84.506 

82.838 
83.671 
84.512 

.99993 
.99993 
.99993 

.21924 
.22359 
.22793 

.91820 
.92254 
.92689 

.91823 
.92257 
.92692 

.99997 
.99997 
.99997 

5.14 
5.15 
5.16 

170.72 
172.43 
174.16 

.00586 
.00580 
.00574 

85.355 
86.213 
87.079 

85.361 
86.219 
87.085 

.99993 
.99993 
.99993 

.23227 
.23662 
.24096 

.93123 
.93557 
.93992 

.93126 
.93560 
.93994 

.99997 
.99997 
.99997 

5.17 
5.18 
5.19 

175.91 
177.68 
179.47 

.00568 
.00563 
.00557 

87.955 
88.839 
89.732 

87.960 
88.844 
89.737 

.99994 
.99994 
.99994 

.24530 
.24965 
.25399 

.94426 
.94860 
.95294 

.94429 
.94863 
.95297 

.99997 
.99997 
.99997 

6.20 

181.27 

.00552 

90.633 

90.639 

.99994 

2.25833 

1.95729 

1.95731 

1.99997 

5.21 
5.22 
5.23 

183.09 
184.93 
186.79 

.00546 
.00541 
.00535 

91.544 
92.464 
93.394 

91.550 
92.470 
93.399 

.99994 
.99994 
.99994 

.26267 
.26702 
.27136 

.96163 
.96597 
.97032 

.96166 
.96600 
.97034 

.99997 
.99997 
.9999e 

5.24 
5.25 
5.26 

188.67 
190.57 
192.48 

.00530 
.00525 
.00520 

94,332 
95.281 
96.238 

94.338 
95.286 
96.243 

.99994 
.99994 
.99995 

.27570 
.28005 
.28439 

.97466 
.97900 
.98335 

.97469 
.97903 
.98337 

.99998 
.99998 
.99998 

5.27 
5.28 
5.29 

194.42 
196.37 
198.34 

.00514 
.00509 
.00504 

97.205 
98.182 
99.169 

97.211 
98.188 
99.174 

.99995 
.99995 
.99995 

.28873 
.29307 
.29742 

.98769 
.99203 
.99638 

.98771 
.99206 
.99640 

.99998 
,99998 
.99998 

6.30 

200.34 

.00499 

100.17 

100.17 

.99995 

2.30176 

2.00072 

2.00074 

1.99998 

5.31 
5.32 
5.33 

202.35 
204.38 
206.44 

.00494 
.00489 
.00484 

101.17 
102.19 
103.22 

101.18 
102.19 
103.22 

.99995 
.99995 
,99995 

.30610 
.31045 
.31479 

.00506 
.00941 
.01375 

.00508 
.00943 
.01377 

.99998 
.99998 
.99998 

5.34 
5.35 
5.36 

208.51 
210.61 
212.72 

.00480 
.00475 
.00470 

104.25 
105.30 
106.36 

104.26 
105.31 
106.36 

.99995 
.99995 
.99996 

.31913 
.32348 
.32782 

.01809 
.02244 
.02678 

.01811 
.02246 
.02680 

.99998 
.99998 
.99998 

5.37 

5.38 
5.39 

214.86 
217.02 
219.20 

.00465 
.00461 
.00456 

107.43 
108.51 
109.60 

107.43 
108.51 
109.60 

.99996 
.99996 
.99996 

.33216 
.33650 
.34085 

.03112 
.03547 
.03981 

.03114 
.03548 
.03983 

.99998 
.99998 
,99998 

6.40 

221.41 

.00452 

110.70 

110.71 

.99996 

2.34519 

2.04415 

2.04417 

1.99998 

1-36 


MATHEMATICS,   UNITS,   AND   SYMBOLS 


X 

_.  Natural  Values 

Common  Logarithms 

«* 

«-* 

Sinhx 

Cosh  a; 

Tanks 

e* 

Sinha; 

Cosh  x 

Tanhx 

5.40 

221.41 

.00452 

110.70 

110.71 

.99996 

2.34519 

2.04415 

2.04417 

1.99998 

5.41 
5.42 
5.43 

223.63 
225.88 
228.15 

.00447 
.00443 
.00438 

111.81 
112.94 
114.07 

111.82 
112.94 
114.08 

.99996 
.99996 
.99996 

.34953 
.35388 
.35822 

.04849 
.05284 
.05718 

.04851 
.05285 
.05720 

.99998 
.99998 
.99998 

5.44 
5.45 
5.46 

230.44 
232.76 
235.10 

.00434 
.00430 
.00425 

115.22 
116.38 
117.55 

115.22 
116.38 
117.55 

.99996 
.99996 
.99996 

.36256 
.36690 
.37125 

.06152 
.06587 
.07021 

.06154 

.06588 
.07023 

.99998 
.99998 
.99998 

5.47 

5.48 
5.49 

237.46 
239.  85 
242.26 

.00421 
.00417 
.00413 

118.73 
119.92 
121.13 

118.73 
119.93 
121.13 

.99996 
.99997 
.99997 

.37559 
.37993 
.38428 

.07455 
.07890 
.08324 

.07457 
.07891 
.08325 

.99998 
.99998 
.99999 

6.60 

244.69 

.00409 

122.34 

122.35 

.99997 

2.38862 

2.08758 

2.08760 

1.99999 

5.51 
5.52 
5.53 

247.15 
249.64 
252.  14 

.00405 
.00401 
.00397 

123.57 
124.82 
126.07 

123.58 
124.82 
126.07 

.99997 
.99997 
.99997 

.39296 
.39731 
.40165 

.09193 
.09627 
.10061 

.09194 
.09628 
.10063 

.99999 
.99999 
.99999 

5.54 
5.55 
5.56 

254.68 
257.24 
259.82 

.00393 
.00389 
.00385 

127.34 
128.62 
129.91 

127.34 
128.62 
129.91 

.99997 
.99997 
.99997 

.40599 
.41033 
.41468 

.10495 
.10930 
.11364 

.10497 
.10931 
.11365 

.99999 
.99999 
.99999 

5.57 
5.58 
5.59 

262.43 
265.07 
267.74 

.00381 
.00377 
.00374 

131.22 
132.53 
133.87 

131.22 
132.54 
133.87 

.99997 
.99997 
.99997 

.41902 
.42336 
.42771 

.11798 
.12233 
.12667 

.11800 
.12234 
.12668 

.99999 
.99999 
.99999 

6.60 

270.43 

.00370 

135.21 

135.22 

.99997 

2.43205 

2.13101 

2.13103 

1.99999 

5.61 
5.62 
5.63 

273.14 
275.89 
278.  66 

.00366 
.00362 
.00359 

136.57 
137.94 
139.33 

136.57 
137.95 
139.33 

.99997 
.99997 
.99997 

.43639 
.44074 
.44508 

.13536 
.13970 
.14404 

.13537 
.13971 
.14405 

.99999 
.99999 
.99999 

5.64 
5.65 
5.66 

281.46 
284.29 
287.15 

.00355 
.00352 
.00348 

140.73 
142.14 
143.57 

140.73 
142.15 
143.58 

.99997 
.99998 
.99998 

.44942 
.45376 
.45811 

.14839 
.15273 
.15707 

.14840 
.15274 
.15708 

.99999 
.99999 
.99999 

5.67 

5.68 
5.69 

290.03 
292.  95 
295.  89 

.00345 
.00341 
,00338 

145.02 
146.47 
147.95 

145.02 
146.48 
147.95 

.99998 
.99998 
.99998 

,46245 
.46679 
.47114 

.16141 
.16576 
.17010 

.16142 
.16577 
.17011 

.99999 
.99999 
.99999 

6.70 

298.87 

.00335 

149.43 

149.44 

.99998 

2.47548 

2.17444 

2.17445 

1.99999 

5.71 
5.72 
5.73 

301.87 
304.90 
307.97 

.00331 
.00328 
.00325 

150.93 
152.45 
153.98 

150.94 
152.45 
153.99 

.99998 
.99998 
.99998 

.47982 
.48416 
.48851 

.17879 
.18313 
.18747 

.17880 
.18314 
.18748 

.99999 
.99999 
.99999 

5.74 
5.75 
5.76 

311.06 
314.19 
317.35 

.00321 
.00318 
.00315 

155.53 
157.09 
158.67 

155.53 
157.10 
158.68 

.99998 
.99998 
.99998 

.49285 
.49719 
.50154 

.19182 
.19616 
.20050 

.19182 
.19617 
.20051 

.99999 
.99999 
.99999 

5.77 
5.78 
5.79 

320.54 
323.76 
327.01 

.00312 
.00309 
.00306 

160.27 
161.88 
163.51 

160.27 
161.88 
163.51 

.99998 
.99998 
.99998 

.50588 
.51022 
.51457 

.20484 
.20919 
.21353 

.20485 
.20920 
.21354 

.99999 
.99999 
.99999 

6.80 

330.30 

.00303 

165.15 

165.15 

.99998 

2.51891 

2.21787 

2.21788 

I.  99999 

5.81 
5.82 
5.83 

333.62 
336.97 
340.36 

.00300 
.00297 
.00294 

166.81 
168.48 
170.18 

166.81 
168.49 
170.18 

.99998 
.99998 
.99998 

.52325 
.52759 
.53194 

.22222 
.22656 
.23090 

.22222 
.22657 
.23091 

.99999 
.99999 
.99999 

5.84 
5.85 
5.86 

343.78 
347.23 
350.72 

.00291 
.00288 
.00285 

171.89 
173.62 
175.36 

171.89 
173.62 
175.36 

.99998 
.99998 
.99998 

.53628 
.54062 
.54497 

.23525 
.23959 
.24393 

.23525 
.23960 
.24394 

.99999 
.99999 
.99999 

5.87 
5.88 
5.89 

354.25 
357.81 
361.41 

.00282 
.00279 
.00277 

177.12 
178.90 
180.70 

177.13 
178.91 
180.70 

.99998 
.99998 
.99998 

.54931 
.55365 
.55799 

.24828 
.25262 
.25696 

.24828 
.  25262 
.25697 

.99999 
.99999 
.99999 

6.90 

366.04 

.00274 

182.52 

182.52 

.99998 

2.56234 

2.26130 

2.26131 

1.99999 

5.91 
5.92 
5.93 

368.71 
372.41 
376.15 

.00271 
.00269 
.00266 

184.35 
186.20 
188.08 

184.35 
186.21 
188.08 

.99999 
.99999 
.99999 

.56668 
.57102 
.57537 

.26565 
.26999 
.27433 

.26565 
.27000 
.  27434 

.99999 
.99999 
.99999 

5.94 
1  5.95 
5.96 

379.93 
383.75 
387.  61 

.00263 
.00261 
.00258 

189.97 
191.88 
193.80 

189.97 
191.88 
193.81 

.99999 
.99999 
.99999 

.57971 
.58405 
.58840 

.27868 
.28302 
.28736 

.  27868 
.  28303 
.  28737 

.99999 
.99999 
.99999 

5.97 

5.98 
5.99 

391.51 
395.44 
399.41 

.00255 
.00253 
.00250 

195.75 
197.72 
199.71 

195.75 
197,72 
199.71 

.99999 
.99999 
.99999 

.59274 
.59708 
.60142 

.29171 
.29605 
.30039 

.29171 
.29605 
.30040 

.99999 
.99999 
.99999 

6.00 

403.43 

.00248 

201.71 

201.72 

.99999 

2.60577 

2.30473 

2.30474 

.99999 

TRANSMISSION   UNIT  AND   POWER  REFERENCE   LEVELS      1-37 


14.  BESSEL  FUNCTIONS 

The  chart  below  shows  approximate  values  for  some  representative  Bessel  functions  of 
the  first  kind.  Values  for  higher-order  Bessel  functions  can  be  computed  by  successive 
application  of  the  recurrence  formula 


starting  with  the  values  of  JQ(X)  and  J\(x). 
1.0 


-0.5 


FIG.  1.     Chart  of  Bessel  Functions. 

16.  TRANSMISSION  UNIT  AND  POWER  REFERENCE 
LEVELS.— DECIBELS 

Power  losses  occur  in  all  parts  of  an  electric  circuit;  in  many  circuits,  which  are  built 
up  of  a  number  of  components,  the  easiest  method  of  predicting  overall  efficiencies  is  to 
determine  individual  efficiencies  and  combine  them.  When  amplifiers  are  used,  power 
gains  exist  (as  far  as  the  alternating  signal  current  is  concerned)  and  must  be  considered. 
For,  although  the  useful  power  is  less  than  the  total  power  input,  the  output  signal  power 
may  be  greater  than  the  input  signal  power,  so  that,  using  the  conventional  definition  of 
efficiency,  values  greater  than  100  per  cent  can  be  obtained. 

THE  DECIBEL  AND  THE  NEPER.  In  order  to  avoid  the  multiplication  of  the  indi- 
vidual efficiencies  recourse  has  been  had  to  logarithms  of  the  efficiencies,  giving  measures 
of  efficiency  which  can  be  added  and  subtracted  directly. 

Many  units  have  been  proposed  and  several  have  at  various  times  been  adopted  in  dif- 
ferent localities.  At  the  present  time  two  such  units  are  in  general  use  in  Europe  and  this 
country,  one  based  on  the  napierian  system  of  logarithms,  the  other  based  on  the  decimal 
system  of  logarithms.  The  International  Advisory  Committee  on  Long  Distance  Telephony 
of  Europe  has  recommended  that  both  these  units  be  standardized  and  defined  as  follows: 

The  unit  of  transmission  expresses  the  ratio  of  apparent  or  real  power  in  transmission 
systems.  In  practice,  the  number  of  units  of  transmission  in  a  given  case  is  expressed  in 
terms  of  a  logarithm. 

In  the  case  of  two  powers  Pi  and  Pz  the  number  of  units  is : 

•p 
in  the  napierian  system,  1/2  loge  — 


in  the  decimal  system, 


10810 


The  napierian  unit  is  called  the  neper.     The  decimal  unit  is  called  the  bel. 
submultiple  of  these  units  may  be  used,  as  decineper  and  decibel. 


A  decimal 


1-38  MATHEMATICS,   UNITS,  AND   SYMBOLS 

The  unit  generally  used  in  this  country  is  the  decibel,  which  is  exactly  equivalent  to 
and  was  first  standardized  as  the  "transmission  unit";  it  is  also  exactly  equivalent  to 
the  "sensation  unit"  used  in  acoustic  work.  The  decibel  is  abbreviated  db,  the  trans- 
mission unit  TU,  and  the  sensation  unit  SU.  The  neper  is  called  the  (31  unit.  The  com- 
parative sizes  of  the  two  units  is  given  by  the  fact  that 

logc  ^  =  2.3026  logio  p^  , 

so  that  0.8686  neper  is  equal  to  1  bel.    Also  1  neper  =  11.51  decibels. 

From  the  above  definition  the  number  of  decibels  which  expresses  the  ratio  between 
any  two  powers  is 

N  =  10  logio  ™ 

The  quantity  N  is  called  the  transmission  equivalent  of  the  element  considered.  It  may 
be  readily  evaluated  for  a  particular  ratio  by  multiplying  the  common  logarithm  of  this 
ratio  by  10.  If  Pi  represents  the  delivered  power  and  p2  the  input  power,  N  will  be  nega- 
tive for  power  losses  and  positive  for  power  gains,  since  the  logarithms  of  numbers  less 
than  unity  are  negative. 

The  use  of  the  decibel  may  be  seen  from  the  following.    If  two  circuit  elements,  with  a 

•n  Tp 

ratio  of  power  output  to  power  input  of  ~  and  ~ ,  respectively,  are  connected  in  series, 
the  power  ratio  of  the  combination  is 

Pout  _  PI       Pa 

Pin    ~  P2       P4  ! 

where  NI  and  N2  are  the  transmission  equivalents  of  the  first  and  second  elements,  respec- 
tively. Taking  the  logarithms  of  both  sides  and  multiplying  through  by  10, 

10  log  -rr— —  =  NI  -}-  Nz  —  NT 
•P  in 

It  is  thus  seen  that  any  number  of  transmission  equivalents  can  be  added  (losses  with 
their  associated  minus  sign)  to  obtain  the  transmission  equivalent  of  a  complete  circuit. 

In  making  measurements  of  circuit  efficiency  the  current  ratio,  or  the  voltage  ratio, 
is  usually  more  readily  obtainable  than  the  power  ratio.  Either  of  these  ratios  may  bo 
used  to  specify  the  efficiency  of  the  circuit  when  conditions  are  such  that  it  is  the  square 
root  of  the  power  ratio. 

In  this  case  Ji2  _  Pi 

and  taking  the  square  root  of  both  sides 


By  the  method  used  above  in  the  case  of  power  ratios,  the  transmission  equivalent  is 

N  =  20  logio  7-1  =  20  logic,  ~ 
1  z  J&z  ~ 

It  must  be  remembered  that  this  is  true  only  when  the  current  or  voltage  ratio  is  the  square  root 
of  the  power  ratio,  the  simplest  case  being  where  the  currents  through,  or  voltages  across, 
equal  impedances  are  measured. 

LOGARITHMIC  VOLTAGE  RATIO.  In  measuring  electron-tube  amplifiers,  it  is  fre- 
quently useful  to  measure  the  voltage  gain  of  each  stage  and  compare  the  sum  with  the 
overall  gain  of  the  complete  amplifier.  Such  measurements  are  in  many  cases  conveniently 
made  in  terms  of  voltage  and  are  made  only  with  great  difficulty  in  terms  of  power.  The 
habit  has  grown  up  of  using  the  advantages  of  logarithmic  addition  and  calibrating  ampli- 
fiers in  terms  of  comparison  voltages  without  any  regard  for  the  impedance  relations. 
Furthermore  in  some  instances  the  overall  sensitivity  of  amplifiers  and  complete  radio 
sets  has  frequently  been  expressed  in  terms  of  "decibels  below  1  volt"  with  no  thought  of 
impedance.  According  to  the  above  discussion  this,  of  course,  is  a  misuse  of  the  term. 
Since  it  is  so  convenient,  however,  it  is  a  practice  which  is  likely  to  continue.  Confusion 
can  be  avoided  by  the  association  of  a  new  term  to  this  measurement.  It  has  been  sug- 
gested that  the  abbreviation  dbv  be  used  to  indicate  that  the  logarithmic  ratios  are  in 
terms  of  volts  and  not  in  terms  of  power.  It  has  furthermore  been  suggested  that  the 
dbv  also  carry  the  implication  where  appropriate  that  it  is  below  the  level  of  1  volt.  A 


TRANSMISSION   UNIT  AND   POWER  REFERENCE   LEVELS      1-39 


Decibels  Versus  Power,  Voltage,  and  Current  Ratios 


db 

Current  and 
Voltage  Ratio 

Power  Ratio 

db 

Current  and 
Voltage  Ratio 

Power  Ratio 

Gain 

Loss 

Gain 

Loss 

Gain 

Loss 

Gain 

Loss 

0.1 

.012 

0.9886 

.023 

0.9772 

5.6 

1.905 

0.5248 

3.631 

0.2754 

0.2 

.023 

.9772 

.047 

.9550 

5.7 

1.928 

.5188 

3.715 

.2692 

0.3 

.035 

.9661 

.072 

.9333 

5.8 

1.950 

.5129 

3.802 

.2630 

0.4 

.047 

.9550 

.097 

.9120 

5.9 

1.973 

.5070 

3.891 

.2570 

0.5 

.059 

.9441 

.122 

.8913 

6.0 

1.995 

.5012 

3.981 

.2512 

0.6 

.072 

.9333 

.148 

.8710 

6.1 

2.018 

.4958 

4.074 

.2455 

0.7 

.084 

.9226 

.175 

.8511 

6.2 

2.042 

.4898 

4.169 

.2399 

0.8 

.097 

.9120 

.202 

.8318 

6.3 

2.065 

.4842 

4.266 

.2344 

0.9 

.109 

.9016 

.230 

.8128 

6.4 

2.089 

.4786 

4.365 

.2291 

1.0 

.122 

.8913 

.259 

.7943 

6.5 

2.114 

.4732 

4.467 

.2239 

.1 

.135 

.8811 

.288 

.7763 

6.6 

2.138 

.4677 

4.571 

.2188 

.2 

.148 

.8710 

.318 

.7586 

6.7 

2.163 

.4624 

4.677 

.2138 

.3 

.162 

.8610 

.349 

.7413 

6.8 

2.188 

.4571 

4.786 

.2089 

.4 

.175 

.8511 

.380 

.7244 

6.9 

2.213 

.4519 

4.898 

.2042 

'    .5 

.189 

.8414 

.413 

.7080 

7.0 

2.239 

.4467 

5.012 

.1995 

.6 

.202 

.8318 

.445 

.6918 

7.1 

2.265 

.4416 

5.129 

.1950 

.7 

.216 

.8222 

.479 

.6761 

7.2 

2.291 

.4365 

5.248 

.1906 

.8 

.230 

.8128 

.514 

.6607 

7.3 

2.317 

.4315 

5.370 

.1862 

.9 

.245 

.8035 

.549 

.6457 

7.4 

2.344 

.4266 

5.495 

.1820 

2.0 

.259 

.7943 

.585 

.6310 

7.5 

2.371 

.4217 

5.623 

.1778 

2.1 

.274 

.7852 

.622 

.6166 

7.6 

2.399 

.4169 

5.754 

.1738 

2.2 

.288 

.7763 

.660 

.6026 

7.7 

2.427 

.4121 

5.888 

.1698 

2.3 

.303 

.7674 

.698 

.5888 

7.8 

2.455 

.4074 

6.026 

.1660 

2.4 

.318 

.7586 

.738 

.5754 

7.9 

2.483 

.4027 

6.  166 

.1622 

2.5 

.334 

.7499 

.778 

.5623 

8.0 

2.512 

.3981 

6.310 

.1585 

2.6 

.349 

.7413 

.820 

.5495 

8.1 

2.541 

.3936 

6.457 

.1549 

2.7 

.365 

.7328 

.862 

.5370 

8.2 

2.570 

.3891 

6.607 

.1514 

2.8 

.380 

.7244 

.905 

.5248 

8.3 

2.600 

.3846 

6.761 

.1479 

2.9 

.396 

.7161 

1.950 

.5129 

8.4 

2.630 

.3802 

6.918 

.1445 

3.0 

.413 

.7080 

1.995 

.50'!  2 

8.5 

2.661 

.3758 

7.079 

.1413 

3.1 

.429 

.6998 

2.042 

.4898 

8.6 

2.692 

.3715 

7.244 

.1380 

3.2 

.445 

.6918 

2.089 

.4786 

8.7 

2.723 

.3673 

7.413 

.1349 

3.3 

.462 

.6839 

2.138 

.4677 

8.8 

2.754 

.3631 

7.586 

.1318 

3.4 

.479 

.6761 

2.188 

.4571 

8.9 

2.786 

.3589 

7.762 

.1288 

3.5 

.496 

.6683 

2.239 

.4467 

9.0 

2.818 

.3548 

7.943 

.1259 

3.6 

.514 

.6607 

2.291 

.4365 

9.1 

2.851 

.3508 

8.128 

.1230 

3.7 

.531 

.6531 

2.344 

.4266 

9.2 

2.884 

.3467 

8.318 

.1202 

3.8 

.549 

.6457 

2.399 

.4169 

9.3 

2.917 

.3428 

8.511 

.1175 

3.9 

.567 

.6383 

2.455 

.4074 

9.4 

2.951 

.3389 

8.710 

.1148 

4.0 

.585 

.6310 

2.512 

.3981 

9.5 

2.985 

.3350 

8.913 

.1122 

4.1 

.603 

.6237 

2.570 

.3891 

9.6 

3.020 

.3311 

9.120 

.1097 

4.2 

.622 

.6166 

2.630 

.3802 

9.7 

3.055 

.3273 

9.333 

.1072 

4.3 

.641 

.6095 

2.692 

.3715 

9.8 

3.090 

.3236 

9.550 

.1047 

4.4 

.660 

.6026 

2.754 

.3631 

9.9 

3.126 

.3199 

9.772 

.1023 

4.5 

.679 

.5957 

2*818 

.3548 

10.0 

3.162 

.3162 

10.000 

.1000 

4.6 

.698 

.5888 

2.884 

.3467 

10.1 

3.199 

.3126 

10.23 

.0977 

4.7 

.718 

.5821 

2.951 

.3389 

10.2 

3.236 

,3090 

10.47 

.0955 

4.8 

.738 

,5754 

3.020 

.3311 

10.3 

3.273 

.3055 

10.72 

.0933 

4.9 

.758 

.5689 

3.090 

.3236 

10.4 

3.311 

.3020 

10.97 

.0912 

5.0 

.778 

.5623 

3.162 

.3162 

10.5 

3.350 

.2985 

11.22 

.0891 

5.1 

.799 

.5559 

3.236 

.3090 

10.6 

3.388 

.2951 

11.48 

.0871 

5.2 

.820 

.5495 

3.311 

.3020 

10.7 

3.428 

.2917 

11.75 

.0851 

5.3 

.841 

.5433 

3.388 

.2951 

10.8 

3.467 

.2884 

12.02 

.0832 

5.4 

.862 

.5370 

3.467 

.2884 

10.9 

3.508 

.2851 

12.30 

.0813 

5.5 

.884 

.5309 

3.548 

.2818 

11.0 

3.548 

.2818 

12.59 

.0794 

1-40  MATHEMATICS,  UNITS,  AND  SYMBOLS 

Decibels  Versus  Power,  Voltage,  and  Current  Ratios — Continued 


db 

Current  and 
Voltage  Ratio 

Power  Ratio 

db 

Current  and 
Voltage  Ratio 

Power  Ratio 

Gain 

Loss 

Gain 

Loss 

Gain 

Loss 

Gain 

Loss 

11.1 

3.589 

0.2786 

12.88 

0.0776 

16.1 

6.383 

0.1566 

40.74 

0.0245 

11.2 

3.631 

.2754 

13.18 

.0759 

16.2 

6.457 

.1549 

41.69 

.0239 

11.3 

3.673 

.2723 

13.49 

.0741 

16.3 

6.531 

.1531 

42.66 

.0234 

11.4 

3.715 

.2692 

13.81 

.0724 

16.4 

6.607 

.1514 

43.65 

.0229 

11.5 

3.758 

.2661 

14.13 

.0708 

16.5 

6.683 

.1496 

44.67 

.0224 

11.6 

3.802 

.2630 

14.45 

.0692 

16.6 

6.761 

.1479 

45.71 

.0219 

11.7 

3.846 

.2600 

14.79 

.0676 

16.7 

6.839 

.1462 

46.77 

.0214 

11.8 

3.891 

.2570 

15.14 

.0661 

16.8 

6.918 

.1445 

47.86 

.0209 

11.9 

3.936 

.2541 

15.49 

.0646 

16.9 

6.998 

.1429 

48.98 

.0204 

12.0 

3.981 

.2512 

15.85 

.0631 

17.0 

7.079 

.1413 

50.12 

.0200 

12.1 

4.027 

.2483 

16.22 

.0617 

17.1 

7.161 

.1396 

51.29 

.0195 

12.2 

4.074 

.2455 

16.60 

.0603 

17,2 

7.244 

.1380 

52.43 

.0191 

12.3 

4.121 

.2427 

16.98 

.0589 

17.3 

7.328 

.1365 

53.70 

.0186 

12.4 

4.169 

.2399 

17.38 

.0575 

17.4 

7.413 

.1349 

54.96 

.0182 

12.5 

4.217 

.2371 

17.78 

.0562 

17.5 

7.499 

.1334 

56.23 

.0178 

12.6 

4.266 

.2344 

18.20 

.0550 

17.6 

7.586 

.1318 

57.54 

.0174 

12.7 

4.315 

.2317 

18.62 

.0537 

17.7 

7.674 

.1303 

58.88 

.0170 

12.8 

4.365 

.2291 

19.05 

.0525 

17.8 

7.762 

.1288 

60.26 

.0166 

12.9 

4.416 

.2265 

19.50 

.0513 

17.9 

7.852 

.1273 

61.66 

.0162 

13.0 

4.467 

.2239 

19.95 

.0501 

18.0 

7.943 

.1259 

63.10 

.0158 

13.1 

4.519 

.2213 

20.42 

.0490 

18.1 

8.035 

.1245 

64.57 

.0155 

13.2 

4.571 

.2188 

20.89 

.0479 

18.2 

8.128 

.1230 

66.07 

.0151 

13.3 

4.624 

.2163 

21.38 

.0468 

18.3 

8.222 

.1216 

67.61 

.0148 

13.4 

4.677 

.2138 

21.88 

.0457 

18.4 

8.318 

.1202 

69.18 

.0145 

13.5 

4.732 

.2113 

22.39 

.0447 

18.5 

8.414 

.1189 

70.80 

.0141 

13.6 

4.786 

.2089 

22.91 

.0437 

18.6 

8.511 

.1175 

72.44 

.0138 

13.7 

4.842 

.2065 

23.44 

.0427 

18.7 

8.610 

.1161 

74.13 

.0135 

13.8 

4.898 

.2042 

23.99 

.0417 

18.8 

8.710 

.1148 

75.86 

.0132 

13.9 

4.955 

.2018 

24.55 

.0407 

18.9 

8.811 

.1135 

77.63 

,0129 

14.0 

5.012 

.1995 

25.12 

.0398 

19.0 

8.913 

.1122 

79.43 

.0126 

14.1 

5.070 

.1972 

25.70 

.0389 

19.1 

9.016 

.1109 

81.28 

,0123 

14.2 

5.129 

.1950 

26.30 

.0380 

19.2 

9.120 

.1097 

83.18 

.0120 

14.3 

5.188 

,1928 

26.92 

.0372 

19.3 

9.226 

.1084 

85.11 

.0117 

14.4 

5.248 

.1906 

27.54 

.0363 

19.4 

9.333 

.1072 

87.10 

.0115 

14.5 

5.309 

.1884 

28.18 

.0355 

19.5 

9.441 

.1059 

89,13 

.0112 

14.6 

5.370 

.1862 

28.84 

.0347 

19.6 

9.550 

.1047 

91.20 

.0110 

14.7 

5.433 

.1841 

29.51 

.0339 

19.7 

9.661 

,1035 

93.33 

.0107 

14.8 

5.495 

.1620 

30.20 

.0331 

19.8 

9.772 

.1023 

95.50 

.0105 

14.9 

5.559 

.1799 

30.90 

.0324 

19.9 

9.886 

.1012 

97.72 

,0102 

15.0 

5.623 

.1778 

31.62 

.0316 

20.0 

10.000 

.1000 

100.0 

.0100 

15.1 

5.689 

.1758 

32.36 

.0309 

30.0 

31.62 

.0316 

1,000 

.0010 

15.2 

5.754 

.1738 

33.11 

.0302 

40.0 

100.0 

.0100 

104 

10-4 

15.3 

5.821 

.1718 

33.88 

.0295 

50.0 

316.2 

.0032 

105 

10-5 

15.4 

5.888 

.1698 

34.67 

.0288 

60.0 

1,000.0 

.0010 

108 

10-8 

15.5 

5.957 

.1679 

35.48 

.0282 

70.0 

3f162.*0 

.0003 

107 

10-7 

15.6 

6.026 

.1660 

36.31 

.0275 

80.0 

0,000.0 

.0001 

108 

10-8 

15.7 

6.096 

,1641 

37.15 

.0269 

90.0 

1,620.0 

.00003 

109 

10-9 

15.8 

6.166 

.1622 

38.02 

.0263 

100.0 

00,000,0 

.00001 

15.9 

6.237 

,1603 

38.91 

,0257 

16.0 

6.310 

.1585 

39.81 

.0251 

BIBLIOGRAPHY  1-41 

sensitivity  of  100  AIV  could  be  expressed  as  —80  dbv,  and  a  sensitivity  of  10  iiv  as  — 100 
dbv,  with  this  system. 

A  table  is  appended  giving  values  of  transmission  equivalents  in  terms  of  both  power 
and  current,  or  voltage,  ratios  in  tenths  of  a  decibel  up  to  20  db.  For  values  above  20  db 
the  tables  may  be  used  as  described  below. 

Example.     To  find  the  current  and  power  ratios  for  a  loss  of  57.6  db. 

1.  The  power  ratio  of  50  db  is  10~5  (this  being  the  first  power  ratio  which  is  an  even 
submultiple  of  10  and  corresponds  to  less  than  57.6  db). 

2.  The  power  ratio  of  7.6  (57.6  -  50  =  7.6)  db  is  0.1738. 

3.  To  add  decibels,  power  ratios  must  be  multiplied,  hence: 

Power  ratio  of  57.6  db  =  0.1738  X  10~5 

4.  The  current  ratio  of  40  db  is  0.01  (first  current  ratio  which  is  an  even  submultiple 
of  10  and  corresponds  to  less  than  57.6  db) . 

5.  The  current  ratio  of  17.6  (57.6  -  40  =  17.6)  db  is  0.1318. 

6.  Multiplying  these  ratios: 

Current  ratio  of  57.6  db  =  0.001318 

POWER  REFERENCE  LEVELS.  When  the  efficiency  of  a  device  or  system  is  ex- 
pressed in  decibels  there  is  in  general  no  indication  of  the  actual  amount  of  power  in  the 
device.  In  comparing  devices  it  is  frequently  desirable  to  know  the  actual  overall  effi- 
ciency. In  such  a  case  this  can  readily  be  expressed  in  decibels,  100  per  cent  efficiency 
being  represented  by  zero  decibels.  In  many  cases,  however,  it  is  more  desirable  that  the 
relative  efficiencies  at  different  frequencies  be  known ;  in  some  such  cases  it  is  also  desirable 
that  the  normal  power  capacity  of  the  device  be  specified  in  such  form  as  to  be  readily 
comparable  with  similar  devices. 

For  such  a  specification  to  be  made  when  the  ordinates  of  a  characteristic  are  in  deci- 
bels, it  is  only  necessary  to  specify  some  arbitrary  amount  of  power  as  corresponding  to 
zero  decibels;  then  every  value  of  decibels  represents  a  definite  amount  of  power  (or  vol- 
ume level).  The  amount  of  power  chosen  as  the  reference  level  is  completely  arbitrary; 
hence  it  has  been  customary  to  choose  some  average  value  of  power  as  zero  level,  for  a 
particular  type  of  work. 

Attempts  at  standardization  have  been  made,  with  the  result  that  the  American  Stand- 
ards Association  (and  IRE)  has  recommended  1  milliwatt  in  600  ohms  in  connection  with 
a  particular  meter  to  measure  levels  in  radio  program  transmission  (see  ASA  "American 
Recommended  Practice  for  Volume  Measurements  of  Electrical  Speech  and  Program 
Waves,"  Nov.  6,  1942)  and  has  introduced  the  term  vu  (vee-you)  to  represent  the  number 
of  decibels  above  or  below  this  level.  Also  it  has  become  customary  to  specify  power  in 
dbm  which  is  used  to  mean  decibels  above  or  below  1  milliwatt. 

However,  some  other  groups  are  still  using  other  levels.  For  instance,  in  sound-motion 
pictures  the  reference  level  is  6  milliwatts.  Also  certain  radar  engineers  use  decibels  below 
or  above  1  watt. 

It  will  apparently  require  further  action  by  the  American  Standards  Association  to  bring 
order  out  of  the  present  chaos.  The  desirability  of  such  a  standard  is  shown  by  the  expe- 
rience in  the  acoustic  field  where  10  ~16  watt  per  sq  cm  was  universally  adopted,  so  that 
measurements  made  anywhere  are  everywhere  intelligible.  Until  the  adoption  of  such  a 
standard,  great  care  must  be  exercised  in  comparing  curves  and  statements  of  levels  to 
insure  that  correction  is  made  for  differences  in  reference  levels.  Also  a  statement  of  the 
reference  used  should  always  be  included  as  a  part  of  any  publication  of  results. 

BIBLIOGRAPHY 

Hilliard,  J.  K.,  Definition  of  Standard  Reference  Systems,  Electronics,  Vol.  3,  192  (November  1931). 
Martin,  W.  H.,  Decibel — The  Name  for  the  Transmission  Unit,  B.S.T.J.,  Vol.  8,  1  (January  1929). 
Martin,  W.  H.,  and  C.  H.  G.  Gray.  Master  Reference  System  for  Telephone  Transmission.  B.S.T.J., 
Vol.  8,  536  (July  1929). 


1-42  MATHEMATICS,  UNITS,  AND  SYMBOLS 

UNITS  AND  CONVERSION  FACTORS 

By  J.  G.  Brainerd  and  Carl  C.  Chambers 

16.  SYSTEMS  OF  UNITS 

The  magnitude  of  a  physical  quantity  has  no  tangible  meaning  except  as  the  relative 
magnitude  of  that  quantity  as  compared  with  some  other  quantity  of  the  same  nature. 
Thus,  50  ohms  is  a  resistance  having  a  magnitude  50  times  the  resistance  of  1  ohm.  There- 
fore, whenever  it  is  necessary  or  desirable  to  talk  about  the  magnitude  of  a  physical^ 
quantity,  it  is  necessary  to  have  a  basis  for  comparison.  This  basis  for  a  quantity  is 
called  the  unit  of  magnitude  of  that  quantity.  In  order  to  communicate  the  idea  of 
magnitude  between  different  people,  it  is  necessary  that  they  at  least  know  the  relative 
magnitudes  of  their  units.  It  is  the  purpose  of  this  section  to  act  as  tool  for  the  specifica- 
tion of  the  relative  magnitudes  of  the  more  commonly  used  systems  of  units  for  physical 
quantities. 

Because  of  the  relations  denning  physical  laws,  there  are  relations  between  the  magni- 
tudes of  physical  quantities.  It  is  desirable  that  these  physical  relations  be  expressed 
alike  hi  the  different  systems  of  units.  For  instance,  the  relation  mass  X  acceleration 
=  force  should  be  independent  of  the  system  of  units.  Therefore,  unit  mass  times  unit 
acceleration  should  equal  unit  force.  This  gives  a  relation  among  these  three  units. 

Because  of  such  physical  relations,  all  the  mechanical  units  can  be  derived  from  the 
units  for  three  fundamental  quantities.  The  three  quantities  ordinarily  taken  as  funda- 
mental are  mass,  length,  and  time.  Thermal  quantities  are  conveniently  derived  from 
these  three  quantities  together  with  another  fundamental  quantity,  temperature.  Photo- 
metric quantities  are  derived  from  the  three  fundamental  mechanical  quantities  together 
with  luminous  intensity  as  a  fourth  fundamental  quantity. 

Similarly,  electrical  and  magnetic  quantities  are  derived  from  the  three  fundamental 
mechanical  quantities  and  one  fundamental  electrical  or  magnetic  quantity. 

Two  systems  of  mechanical  units  are  in  use  in  English-speaking  countries,  the  English 
and  the  metric  systems.  The  metric  system  is  used  universally  by  physicists  and  to  a 
great  extent  by  engineers,  although  the  English  system  is  still  very  common  in  engineering. 
The  English  system  uses  the  foot,  the  pound,  and  the  second  as  the  units  for  length,  mass, 
and  time,  respectively.  The  metric  system  (as  used  in  the  current  literature  —  see  MKS 
system  below)  employs  the  meter,  the  kilogram,  and  the  second  as  the  units  for  length, 
mass,  and  time,  respectively. 

STANDARDS  OF  THE  FUNDAMENTAL  UNITS.  The  physical  units  upon  which 
these  fundamental  units  are  based  and  the  legalized  standards  of  the  United  States  and 
Great  Britain  are  described  below. 

Standard  of  Length.  The  standard  meter  (100  cm)  is  the  distance  between  two  lines 
on  a  platinum-iridium  bar  carefully  preserved  at  the  Bureau  of  Weights  and  Measures, 
at  Sevres,  France,  when  the  bar  is  kept  at  a  uniform  temperature  of  0  deg  cent  throughout. 
In  the  United  States  the  yard  (3  ft)  was  defined  by  Act  of  Congress,  July  28,  1866,  as 

1  U.  S.  yard  =  7        meter 


and  similarly  the  British  imperial  yard  is  defined  by  law  as 

1  British  imperial  yard  «  3Qg7  Q7Q  meter 

For  engineering  purposes  the  U.  S.  and  British  yards  may  be  considered  identical. 

Standard  of  Mass  and  Force.  The  standard  kilogram  (1000  grams)  is  the  mass  of 
a  cylinder  of  platinum  preserved  at  the  Bureau  of  Weights  and  Measures,  at  Sevres 
France.  The  U.  S.  pound  avoirdupois  is  defined  by  law  (Act  of  Congress,  1866)  as 

2.2046  kg'  but  k  1S93'  the  SuPerinten<knt  of  Weights  and  Measures,  with  the  approval  of 
the  Secretary  of  the  Treasury,  declared  the  pound  to  be 

1U'S-lb  =  2 
The  British  imperial  pound  has  the  same  value. 

The  same  relations  between  the  pound  and  kilogram  hold  whether  these  units  be  taken 
as  units  of  mass  or  as  units  of  force,  the  unit  of  force  being  defined  in  both  cases  as  the  pull 
of  the  earth  on  unit  mass  at  45  deg  latitude  and  sea  level 


SYSTEMS  OF  UNITS  1-43 

Standard  of  Time.     The  standard  second  universally  adopted  is  the  -zrr-:-:^  part  of  a 


mean  solar  day.  The  solar  day  is  the  interval  of  time  between  two  successive  transits  of 
the  sun  across  a  meridian  of  the  earth  at  the  point  of  observation;  this  interval  varies 
in  length  at  different  times  during  the  year,  but  the  average  length  of  the  interval  for  one 
year  is  constant  as  far  as  can  be  determined  by  any  known  methods  of  observation. 

Temperature  Scales.  Two  units  of  temperature,  or  temperature  scales,  are  commonly 
employed,  viz.,  the  centigrade  and  the  fahrenheit  units.  The  relation  between  these  two 
units  results  solely  from  the  manner  in  which  they  are  denned.  One  degree  centigrade 
=  %  degree  fahrenheit.  Owing  to  the  difference  in  the  zeros  of  the  two  scales,  a  tem- 
perature of  tf  degrees  fahrenheit  corresponds  to  a  temperature  of  tc  =  5/9(£/  —  32)  degrees 
centigrade,  and  vice  versa,  t/  —  9/5  tc  +  32  degrees  fahrenheit. 

Standard  of  Luminous  Intensity.  Before  Jan.  1,  1948,  the  standard  of  luminous  in- 
tensity was  the  mean  intensity  in  the  horizontal  plane  from  a  group  of  incandescent  lamps 
maintained  by  the  National  Bureau  of  Standards  (U.  S.),  in  cooperation  with  similar 
custodians  in  France,  Great  Britain,  and  Germany.  The  International  candle  was  a  point 
source  of  light  having  an  intensity  of  a  definite  fraction  of  this  standard  intensity. 

The  National  Bureau  of  Standards,  in  pursuance  of  decisions  of  the  International  Com- 
mittee on  Weights  and  Measures,  decided  that,  beginning  Jan.  1,  1948,  it  would  take  as 
the  primary  standard  for  the  system  of  photometric  units  a  black-body  radiator  operated 
at  the  temperature  of  freezing  platinum.  The  "candle,"  unit  of  intensity,  is  denned  as 
one-sixtieth  of  the  intensity  of  one  square  centimeter  of  such  a  radiator.  Other  units  are 
derived  from  the  candle,  with  the  provision  that  when  differences  of  color  are  involved  the 
evaluation  shall  be  made  by  means  of  standard  spectral  luminosity  factors  which  have  been 
adopted  by  the  International  Commission  on  Illumination  and  the  International  Com- 
mittee on  Weights  and  Measures. 

ELECTRIC  UNITS.  Three  systems  of  electric  and  magnetic  units  are  in  general  use, 
viz.,  (1)  the  cgs  electrostatic  system,  (2)  the  cgs  electromagnetic  system,  and  (3)  the 
practical  system.  In  the  cgs  electrostatic  system  the  dielectric  coefficient,  «,  of  air  *  at 
0  deg  cent  and  760  mm  mercury  pressure  is  arbitrarily  chosen  as  unity.  In  the  cgs  elec- 
tromagnetic system  the  magnetic  permeability  of  air  under  the  same  standard  conditions 
is  arbitrarily  chosen  as  unity.  In  the  practical  system  a  concrete  standard  of  the  unit  of 
resistance  (called  the  ohm)  and  of  the  unit  of  current  (ampere)  is  arbitrarily  chosen 
(it  was  stated  above  that  only  one  electric  or  magnetic  unit  need  be  chosen;  the  choice  of 
two  leads  to  inconsistencies;  see  below)  ;  the  unit  of  resistance  is  closely  equal  to  109  times 
the  unit  of  resistance  in  the  cgs  electromagnetic  system  and  the  unit  current  is  approxi- 
mately 0.1  that  in  the  latter  system.  Occasionally  other  (special)  systems  are  used,  most 
of  which  are  designed  to  get  rid  of  a  factor  4?r  which  frequently  appears  in  the  usual 
systems.  The  most  popular  of  these  others  is  the  Heaviside—  Lorentz  system  in  which 
the  unit  of  electric  charge  is  I/V^TT  of  the  unit  in  the  electrostatic  system.  (See  MKS 
system.) 

Use  of  the  Prefixes  "Stat"  and  "Ab."  To  designate  the  electric  and  magnetic  units 
in  the  electrostatic  and  electromagnetic  systems  of  units  respectively,  the  prefixes  "stat" 
and  "ab"  may  be  used  with  the  name  of  the  corresponding  practical  unit.  For  example, 
the  cgs  electrostatic  unit  of  electric  charge  may  be  called  the  statcoulomb  and  the  cgs 
electromagnetic  unit  of  electric  charge  may  be  called  the  abcoulomb,  etc.  f 

Relations  among  the  Three  Systems  of  Electrical  Units.  The  fundamental  relations, 
experimentally  determined,  between  the  cgs  electrostatic  and  the  cgs  electromagnetic 
system  is  that  1  abfarad  —  8.9878  X  1020  statfarads,  which  may  be  approximated  for 
engineering  purposes  to 

1  abfarad  =  9  X  1020  statfarads 

which,  as  a  consequence  of  the  definition  of  the  various  terms,  is  equivalent  to 

1  abcoulomb  =  3  X  1010  stat  coulombs 
the  erg  being  the  unit  of  energy  in  both  systems.    Rigorously, 

1  abcoulomb  -  2.9979  X  1010  statcoulombs 
(See  the  article  by  Birge,  Rev.  of  Mod.  Pkys.,  Vol.  1,  1  [July  1929].) 

*  Rigorously,  eo  of  free  or  empty  space  is  chosen  unity;  for  air  at  0  deg  cent  and  76  cm  mercury 
pressure  <=o  =  1.000585;  see  International  Critical  Tables,  Vol.  6,  77,  for  the  value  of  «  for  air  under 
various  conditions. 

t  This  abcoulomb,  the  unit  of  quantity  of  electricity  in  the  electromagnetic  system,  should  not  be 
confused  with  an  "absolute  coulomb,"  which  is  a  unit  closely  equal  to  the  coulomb  and  is  what  the 
latter  would  be  if  1  international  or  practical  ohm  equaled  exactly  109  abohms  and  1  ampere  equaled 
exactly  0.1  abamp.  For  engineering  purposes,  the  difference  between  an  absolute  coulomb  and  a 
coulomb  is  negligible. 


1_44  MATHEMATICS,  UNITS,   AND   SYMBOLS 

The  fundamental  relations  between  the  cgs  electromagnetic  system  and  the  practical 
system  are 

1  abcoulomb  =10  coulombs 

1  erg  =  10 ~7  watt-second  or  joule 

the  erg  being  the  unit  of  energy  in  the  cgs  electromagnetic  system  and  the  joule  (or  watt- 
second)  that  in  the  practical  system. 

Practical  Electrical  Units.  The  former  (see  below)  legal  units  of  electrical  measure  in 
the  United  States  are  given  in  an  Act  of  Congress,  July  12,  1894.  Unfortunately,  the 
units  there  defined  are  not  consistent  with  one  another;  for  example,  the  unit  of  power 
(watt)  there  given  is  not  equal  to  the  unit  of  power  derived  from  the  units  of  current 
(ampere)  and  voltage  (volt)  as  defined  in  the  Act.  The  practical  units  (the  so-called 
international  units)  in  use  before  Jan.  1,  1948,  are  based  on  the  following  two  definitions: 

The  unit  of  resistance  is  the  (international)  ohm  and  is  equal  to  the  resistance  offered 
to  an  unvarying  electric  current  by  a  column  of  mercury  at  the  temperature  of  melting  ice, 
14.4521  grams  in  mass,  of  a  constant  cross-sectional  area  and  106.300  cm  in  length. 

The  unit  of  current  is  the  (international)  ampere  and  is  equal  to  the  unvarying  electric 
current  which,  when  passed  through  a  solution  of  nitrate  of  silver  in  accordance  with 
certain  specifications,  deposits  silver  at  the  rate  of  0.00111800  gram  per  second. 

The  unit  of  electromotive  force,  the  (international)  volt,  is  derived  from  the  above  by 
Ohm's  law.  Other  international  units  are  derived  from  these. 

The  National  Bureau  of  Standards,  in  agreement  with  decisions  of  the  International 
Committee  on  Weights  and  Measures,  decided  to  use  as  standard,  beginning  Jan.  1,  1948, 
the  electrical  units  "derived  from  the  fundamental  mechanical  units  of  length,  mass,  and 
time  by  use  of  accepted  principles  of  electromagnetism,  with  the  value  of  the  permeability 
of  space  taken  as  unity  in  the  eentimeter-gram-second  system  or  as  10  ~7  in  the  correspond- 
ing meter-kilogram-second  system."  The  reference  is  to  the  unrationalized  MKS  sys- 
tem; in  the  rationalized  MKS  system,  the  permeability  of  space  is  4n-  X  10~7  henry  per 
meter. 

In  explanation  of  the  legal  status  of  the  new  standard,  the  Bureau  states,  "When  the 
electrical  units  were  defined  by  law  (Public  Law  No.  105,  53rd  Congress)  in  1894  it  was 
supposed  that  the  international  units  were  practically  identical  with  the  corresponding 
multiples  of  the  centimeter-gram-second  electromagnetic  system.  Alternative  definitions 
were  given  for  most  of  the  units,  and  those  definitions  which  appear  to  be  legally  control- 
ling were  taken  partly  from  one  system  and  partly  from  the  other.  The  joule  and  the 
watt,  for  example,  are  clearly  denned  as  multiples  of  the  cgs  units.  In  brief,  the  absolute 
units  have  as  good  a  legal  basis  under  the  terms  of  that  act  as  do  the  present  international 
units.  New  legislation  is  being  proposed  to  remove  the  ambiguities  of  the  old  act,  but 
there  should  be  no  objection  on  legal  grounds  to  the  general  adoption  of  the  absolute  units 
even  in  advance  of  Congressional  action." 

Using  "international"  to  refer  to  the  previous  standard,  and  "absolute"  to  refer  to  the 
new,  the  relations  accepted  by  the  International  Committee  on  Weights  and  Measures  at 
its  meeting  in  Paris  in  October,  1946,  are  as  follows: 

1  mean  international  ohm  =  1.00049  absolute  ohms 
1  mean  international  volt  =  1.00034  absolute  volts 

The  mean  international  units  to  which  the  above  equations  refer  are  the  averages  of 
units  as  maintained  in  the  national  laboratories  of  the  six  countries  (France,  Germany, 
Great  Britain,  Japan,  U.S.S.R.,  and  the  United  States)  which  took  part  in  this  work  be- 
fore the  war.  The  units  maintained  by  the  National  Bureau  of  Standards  differ  from 
these  average  units  by  a  few  parts  in  a  million,  so  that  the  conversion  factors  for  adjusting 
values  of  standards  in  this  country  will  be  as  follows  : 

1  international  ohm  (U.  S.)  =  1.000495  absolute  ohms 
1  international  volt  (U.  S.)  =  1.00033  absolute  volts. 
Other  electrical  units  will  be  changed  by  amounts  shown  in  the  following  table: 

1  international  ampere  =  0.999835  absolute  ampere 

1  international  coulomb  «  0.999835  absolute  coulomb 

1  international  henry  =  1.000495  absolute  henrys 

1  international  farad  =  0.999505  absolute  farad 

1  international  watt  =  1.000165  absolute  watts 

1  international  joule  =  1.000165  absolute  joules 


SYSTEMS   OF  UNITS      *  1-45 

The  Act  of  1894  defined  the  international  ohm  as  previously  stated,  but  denned  the  ampere 
as  0.1  abampere.  These  units  give  rise  to  the  so-called  "semiabsolute"  system,  which  is 
seldom  used. 

THE  MKS  SYSTEM  OF  UNITS.  In  1904  Giorgi  proposed  a  system  of  units  in  which 
the  fundamental  units  were  the  meter,  the  kilogram,  the  second,  and  the  ohm.  Using 
this  system  of  fundamental  units,  the  permeability  of  free  space  is  MO  =  4?r  X  10~7  henry 
per  meter,  and  the  equations  of  electricity  and  magnetism,  using  the  practical  units, 
become  equations  without  factors  such  as  108,  etc.  Such  a  system  is  similar  to  the  so- 
called  absolute  systems  such  as  the  cgs  electromagnetic  and  the  cgs  electrostatic  systems. 
It  follows  from  the  theory  of  radiation  of  electromagnetic  waves  that  the  dielectric  coeffi- 
cient <=o  =  — ;  ,  where  c  is  the  ratio  of  electromagnetic  to  electrostatic  units,  which  can  be 
Moc2 

taken  as  the  velocity  of  light  in  free  space. 

The  International  Committee  of  Weights  and  Measures,  at  its  meeting  in  October  1946, 
decided  that  the  actual  substitution  of  this  absolute  system  of  electrical  units  for  the  inter- 
national system  should  take  place  on  January  1,  1948. 

The  units  are  then  defined  by  a  set  of  definitions  such  as  follows: 

(a)  Ampere.     The  ampere  is  the  constant  current  which,  maintained  in  two  parallel 
rectilinear  conductors  of  infinite  length  separated  by  a  distance  of  1  meter,  produces 
between  these  conductors  a  force  equal  to  2  X  10  ~7  mks  (meter-kilogram-second)  units 
of  force  per  meter  of  length. 

(b)  Volt.     The  volt  is  the  difference  of  electrical  potential  between  two  points  of  a 
conductor  carrying  a  constant  current  of  1  ampere  when  the  power  dissipated  between 
these  points  is  equal  to  1  mks  unit  of  power  (watt). 

(c)  Coulomb.     The  coulomb  is  the  quantity  of  electricity  transported  each  second  by 
a  current  of  1  ampere. 

(d)  Ohm.     The  ohm  is  the  electrical  resistance  between  two  points  of  a  conductor 
when  a  constant  difference  of  potential  of  1  volt,  applied  between  these  points,  produces 
in  the  conductor  a  current  of  1  ampere,  the  conductor  not  being  the  seat  of  an  electro- 
motive force. 

(e)  Weber.     The  weber  is  the  magnetic  flux  which,  traversing  a  circuit  of  a  single 
turn,  would  produce  an  electromotive  force  of  1  volt,  if  brought  to  zero  in  1  second  with 
uniform  diminution. 

(f)  Henry.     The  henry  is  the  inductance  of  a  closed  circuit  in  which  an  electromotive 
force  of  1  volt  is  produced  when  the  electric  current  traversing  the  circuit  varies  uni- 
formly at  the  rate  of  1  ampere  per  second. 

(g)  Farad.     The  farad  is  the  electrical  capacitance  of  a  capacitor  between  the  plates 
of  which  appears  an  electrical  difference  of  potential  of  1  volt,  when  charged  with  1  coulomb 
of  electric  charge. 

The  original  Giorgi  MKS  system  chose  the  ohm  as  the  fourth  fundamental  unit.  This 
choice  has  not  been  confirmed.  The  electrical  fundamental  unit  could  be  almost  any  of 
the  electrical  units.  No  particular  unit  has  as  yet  been  chosen  as  fundamental.  The 
preferences  seem  to  be  divided  between  the  ampere,  the  ohm,  the  permeability,  and  the 
coulomb. 

The  original  Giorgi  MKS  system  chose  fj-o  =  4-7rlO~7  henry  per  meter,  the  4?r  factor  caus- 
ing the  electromagnetic  formulas  expressing  rectilinear  symmetry,  such  as  the  Maxwell 
equations,  to  be  free  of  the  factor  4?r,  and  the  electromagnetic  formulas  expressing  circular 
symmetry,  such  as  Coulomb's  law,  to  contain  the  factor  47r.  Such  a  system  is  called  a 
rationalized  system  as  contrasted  with  a  non-rationalized  system,  examples  of  which  are 
the  electromagnetic  and  the  electrostatic  cgs  systems.  The  non-rationalized  MKS  system 
corresponding  to  the  original  Giorgi  system  is  defined  by  the  choice  of  JUQ  ~  10~7.  This 
changes  the  values  of  some  of  the  units  as  shown  in  the  table  below. 


1-46 


MATHEMATICS,  UNITS,   AND  SYMBOLS 


Rationalized  MKS  Units  and  Corresponding  COS  Electromagnetic  Units 

Multiply  mks  units  by  F  to  obtain  cgs  units 


Quantity 

Symbol 

MKS  Unit 

CGS  Unit 

F 

Mechanical 

JjQQOrtil                                         

L 

m 

cm 

10;* 

M 

kg 

g 

103 

Time       

T 

sec 

sec 

.L 

.Area                      »  

s 

sq  m 

sq  cm 

10* 

V 

cu  m  (stere) 

cu  cm 

106 

f 

cycle  per  sec  (hertz) 

cycle  per  sec 

1 

Density      

d 

kg  per  cu  m 

g  per  cu  cm 

10   3 

Velocity             

V 

m  per  sec 

cm  per  sec 

10* 

a 

m  per  sec  per  sec 

cm  per  sec  per  sec 

10* 

Force                  

F 

newton  (j  per  m). 

dyne 

106 

Pressure  

P 

newton  per  sq  m 

dyne  per  sq  cm 

10 

*,0 

radian 

radian 

1 

.Angular  velocity              .  . 

w 

radian  per  sec 

radian  per  sec 

1 

T 

j  per  radian 

dyne  cm 

10I 

^loment  of  inertia       

J 

kg-sq  m 

g-sq  cm 

107 

Energetics 
"Work  or  eneroy        .        ... 

w 

j 

erg 

107 

Volume  energy  or  energy 
density           .        

w 

j  pei*  cu  m 

erg  per  cu  cm 

10_ 

Active  power  

P 

w 

erg  per  sec 

107 

Reactive  power  

Q 

var 

erg  per  sec 

107 

Thermal 
Quantity  of  heat  

Q 

kg  cal 

g  cal 

103 

Temperature 

e 

C  or  K 

C  or  K 

1 

Luminous 
Intensity                            .  . 

I 

candle 

candle 

1 

t 

1 

1 

1  A 

IlTuminatipn      ...      ...... 

E 

lux 

phot 

lo-J 

Brightness 

b 

C8.nrJl6  p<»f  gq  rn 

stilb 

10~4 

Elec'rical 
Electromotive  force         .  .  . 

E 

volt 

abvolt 

108 

Potential  gradient  or  elec- 
tric field,  intensity 

E 

volt  ppr  m 

abvolt  per  cm 

106 

Resistance         

R 

ohm 

abohm 

109 

Resistivity 

p 

ohm-m 

ab  ohm-cm 

1011 

Conductance 

G 

siemens  mho 

abmho 

10~9 

Conductivity         

y 

mho  per  m 

abmho  per  cm 

lo-11 

§uantity  or  displacement  .  . 
urrent  

} 

coulomb 
amp 

abcoulomb 
abamp 

10-1 

10-1 

Electric  flux       .... 

•% 

coulomb 

abcoulomb 

10~1 

Flux  density   

D 

coulomb  per  sq  m 

abcoulomb  per  sq  cm 

o-6 

Current  density 

i 

ampere  per  sq  m 

abampere  per  sq  cm 

io-B 

Capacitance  

c 

farad 

abfarad 

10"~9 

Specific  inductive  capacity. 
Dielectric     coefficient     for 
free  space  or  space  ca- 
pacitivity 

e/eo 

£Q 

numeric 
107/47rc2  —  8  854  X  10~12 

numeric 
4  =  1.113  X  NT21 

1 

Magnetic 
Magnetomotive  force 

$ 

ftTnp-tlTm 

gilbert 

4?r  1  0""1 

Magnetizing  force  or  mag- 
netic field  intensity  
Space  permeability 

H 

MO 

amp-turn  per  m 
47r]0-7  -  ]  257  x  10~6 

oersted 
1 

47T  1  0~3 

Relative  permeability  
Magnetic  flux  

M/MO 
4> 

numeric 

Weber 

numeric 

108 

Flux  density 

B 

ir)4 

Reluctance  

(R 

amp-turn  per  weber 

gauss 

4T  j  Q^S 

Permeance 

(P 

i  n9  //!_.. 

Inductance 

L 

h          ^          ^ 

,  , 

1  fi9 

Pole  strength  

in 

weber 

aonenry 

108  /4_. 

Magnetization 

cr 

e  i/   TT 

i  n4  /A 

Magnetic  moment  

m 

weber-m 

maxwell-cm/Mr 

1010/4* 

Rationalized  MKS  Units  and  Corresponding  Non-rationalized  Units 

Multiply  non-rationalized  mks  units  by  F  to  obtain  rationalized  mks  units 


Quantity 

Symbol 

Name  of  Rationalized  MKS  Units 

F 

Electrical 
Electric  flux  

¥ 
D 

€0 

M  or  5 
H 

fJ-O 

(P 
(R 

m 

m 

B 

coulomb 
coulomb  per  sq  m 
farad  per  m 

amp-turn 
amp-turn  per  m 
henry  per  m 
weber  per  amp-turn 
amp-turn  per  weber 
weber 
weber-m 
weber  per  sq  m 

4* 
4ir 

47T 
1/47T 

r 

47T 

l/4,r 

47T 
4T 
47T 

Flux  density  .  . 

Space  capacitivity 

Magnetic 
Magnetomotive  force  
Magnetizing  force 

Space  permeability  .  . 

Permeance  

Reluctance  

Pole  strength  

Magnetic  moment  

Flux  density  

CONVERSION  TABLES 


1-47 


17.  CONVERSION  TABLES 
Table  1.    Length  [L] 


Multiply 
N.         Number 

\Kof~* 

N^X 
to     XSN 

Obtain   \>C 

±       N 

1 

•g 

1 

J 

Kilometers 

Nautical  miles 

1 

, 

s 

Millimeters  1 

£ 

Centimeters 

1 

30.48 

2.540 

105 

1.853 
X105 

100 

2.540 
X10-3 

1.609 
X105 

0.1 

91.44 

Feet 

3.281 
XI  0-2 

I 

8.333 
XI  0-2 

3281 

6080.27 

3.281 

8.333 
X10-5 

5280 

3.281 
X10-3 

3 

Inches 

0.3937 

12 

1 

3.937 
XIO* 

7.296 
XIO* 

39.37 

0.001 

6.336 
X10* 

3.937 
XI  0-2 

36 

Kilometers 

10-6 

3.048 
X10-4 

2.540 
X10-6 

1 

1.853 

0.001 

2.540 
X10-8 

1.609 

10-6 

9.144 
X1CM 

Nautical  miles 

1.645 
XI  0-4 

0.5396 

1 

5.396 
X10-4 

0.8684 

4.934 

xio-< 

Meters 

0.01 

0.3048 

2.540 
XI  0-2 

1000 

1853 

1 

1609 

0.001 

0.9144 

Mils 

393.7 

1.2 
X104 

1000 

3.937 

xio? 

3.937 
X104 

1 

39.37 

3.6 
X104 

Miles 

6.214 
XlO-e 

1,894 
XI  0-4 

1.578 
X10-5 

0.6214 

1.1516 

6.214 
XI  O-* 

1 

6.214 
X10-7 

5.682 
X10-4 

Millimeters 

10 

304.8 

25.40 

106 

1000 

2.540 
XI  0-2 

1 

914.4 

Yards 

1.094 
XI  0-2 

0.3333 

2.778 
XI  0-2 

1094 

2027 

1.094 

2.778 
X10-5 

1760 

1.094 
X10-3 

1 

Metric  Multiples 

10fl  microns  —  103  millimeters  =  102  centimeters  =  10  decimeters  =  1  meter 
=  10"1  dekameter  —  10~2  hectometer  «  10~3  kilometer  =  10~4  myriameter 
=  10~8  megameter  =  1010  Angstrom  Units. 

Land  Measure 
7.92  inches     =  1  link 

25  links        =  1  rod  =  16.5  feet  —  5.5  yards  (1  rod  =  1  pole  =  1  perch) 
4  rods         =  1  chain  (Gunther's)  =  66  feet  =  22  yards  =  100  links 
10  chains     =  1  furlong  =  660  feet  =  220  yards  =  1000  links  =  40  rods 
8  furlongs  =  1  mile  =  5280  feet  =  1760  yards  =  8000  links  =  320  rods  =  80  chains 

Ropes  and  Cables 
2  yards  =  1  fathom  120  fathoms  =  1  cable's  length 

Nautical  Measure 

6080.27  feet  =  1  knot  «  1  nautical  mile  —  1.15156  statute  miles  * 

3  nautical  miles  =  1  league  (U.  S.)          3  statute  miles  =  1  league  (Gr.  Britain) 

(NOTE. — A  knot,  or  nautical  mile,  is  the  length  of  a  minute  of  longitude  of  the  earth 
at  the  equator  at  sea  level.  The  British  Admiralty  uses  the  round  figure  of  6080  feet. 
The  word  "knot"  is  frequently  used  also  to  denote  "nautical  miles  per  hour.") 


Miscellaneous 


3  inches  =  1  palm 

4  inches  =  1  hand 


9  inches 

2  1/2  feet  -. 


•  1  span 

:  1  military  pace 


1-48  MATHEMATICS,   UNITS,   AND   SYMBOLS 

Table  2.    Area  |Z2J 


Multiply 
Nv        Number 

Obtain       N!^ 

g 

o 

] 

1 
1 
§ 

i 

I 

8 

I 

02 

Square  kilometers 

| 

Square  miles 

1 

tu 

1 

Square  yards 

Acres 

1 

2.296 
XI  0-5 

247.1 

2  471 
X10-1 

640 

2.066 
XIO-4 

Circular  mils 

1 

1.973 
X10& 

1.833 
XI08 

1.273 
X10G 

1.973 

1973 

Square  centimeters 

5.067 
XI  0-e 

1 

929.0 

6.452 

lOio 

104 

2.590 
XI  0*0 

0.01 

8361 

Square  feet 

4.356 

1.076 
X10-3 

1 

6.944 
XI  0-3 

1.076 
XIO? 

10.76 

2.788 
X107 

1.076 

xio-* 

9 

Square  inches 

6,272,640 

7.854 
XI  0-7 

0.1550 

144 

1 

1.550 

1550 

4.015 
X109 

1.550 
X10-3 

1296 

Square  kilometers 

4.047 
X10-3 

10-10 

9.290 
X10-8 

6.452 
XI  O-io 

1 

10-8 

2.590 

10-12 

8.361 
X10-7 

Square  meters 

4047 

0.0001 

9.290 
XI  0-2 

6.452 
XIO-4 

100 

1 

2.590 
X10C 

10-o 

0.8361 

Square  miles 

1.562 
XI  0-3 

3.861 
XI  O-ii 

3.587 
XI  0-8 

0.3861 

3  861 
XI  0-7 

1 

3.861 
XI  0-13 

3.228 
XIO-7 

Square  millimeters 

5.067 
XI  0-4 

100 

9.290 
XIO* 

645.2 

1012 

106 

1 

8.361 
X105 

Square  yards 

4840 

1.196 
XIO-4 

0.1II1 

7.716 
XIO-4 

1.196 
X106 

1.196 

3,098 

1,196 
X10-6 

1 

Land  Measure 
30  1/4  square  yards   =  1  square  rod  =  272  1/4  square  feet 

16  square  rods      =  1  square  chain  =  484  square  yards  =  4356  square  feet 
2  1/2  square  chains  =  1  rood  =  40  square  rods  =  1210  square  yards 
4  roods  =  1  acre  =  10  square  chains  =160  square  rods 

640  acres  =  1  square  mile  =  2560  roods  =  102,400  square  rods 

1  section  of  land  =  1  square  mile;  1  quarter  section  =160  acres 

Architect's  Measure 
100  square  feet  =  1  square 

Circular  Inch  and  Circular  Mil 

A  circular  inch  is  the  area  of  a  circle  1  inch  in  diameter  =  0.7854  square  inch 
1  square  inch    =  1.2732  circular  inches 

A  circular  mil  is  the  area  of  a  circle  1  mil  (or  0.001  inch)  in  diameter  =  0.7854  square  mil 
1  square  mil     =  1.2732  circular  mils 

1  circular  inch  =  105  circular  mils  =  0.7854  X  106  square  mils 
1  square  inch    =  1.2732  X  105  circular  mils  =  106  square  mils 

Metric  Multiples 
1  square  meter  =  1  centiare  =  10~2  are  =  10™4  hectare 

—  10~~6  square  kilometer  =  10~8  square  myriameter 


CONVERSION  TABLES 


1-49 


Table  3.    Volume  [I8] 


Multiply 

\.        Numfrfif 

E 

'^^ 

I 

•8 

I 

J 

2 
-§^ 
Q   S 

1 

1 

B 

?2 

1 

Obtain     ^VN 
^                  ^ 

« 

I 

o 

I 

1 

o 

.2  J 

I-3 

o 

1 

§ 

1 

-2 

I 

Bushels  (dry) 

1 

0.8036 

4.651 

28.38 

2.838 

xio-4 

XIO-2 

Cubic  centimeters 

3.524 

1 

2.832 

16.39 

106 

7.646 

3785 

1000 

473.2 

946.4 

XIO* 

XIO* 

xio5 

Cubic  feet 

1  .2445 

3.531 

1 

5.787 

35.31 

27 

0.1337 

3.531 

1.671 

3.342 

X10-5 

X10-4 

XIO-2 

xio-s 

XIO-2 

Cubic  bches 

2150.4 

6.102 

1728 

1 

6.102 

46,656 

231 

61.02 

28.87 

57.75 

XIO-2 

XltK 

Cubic  meters 

3.524 

10-6 

2.832 

1.639 

1 

0.7646 

3.785 

0.001 

4.732 

9.464 

(steres) 

XIO-2 

XIO-2 

XI  0-5 

X10-3 

xio-*4 

XIO-* 

Cubic  yards 

1.308 

3.704 

2.143 

1.308 

1 

4.951 

1.308 

6.189 

1.238 

XlO-e 

XIO-2 

X10-5 

XI  0-3 

X10-3 

X10-4 

XlO-s 

Gallons  (liquid) 

2.642 

7.481 

4.329 

264.2 

202.0 

1 

0.2642 

0.125 

0.25 

X10-4 

X10-3 

Liters 

35.24 

0.001 

28.32 

1.639 

1000 

764.6 

3.785 

1 

0.4732 

0.9464 

XIO-2 

Pints  (liquid) 

2.113 

59.84 

3.463 

2113 

1616 

8 

2,113 

1 

2 

XI  0-3 

xio-^ 

t 

Quarts  (liquid)  

1.057 

29.92 

1.732 

1057 

807.9 

4 

1.057 

0.5 

1 

XI  0-3 

XIO-2 

Acre-feet:  multiply  number  of  acre-feet  by  4.356  X  104  to  obtain  number  of  cubic  feet; 
multiply  by  3.259  X  105  to  obtain  number  of  gallons. 

Metric  Multiples 

10  milliliters    =  1  centiliter  =  0.338  fluid  ounce 

10  centiliters  =  1  deciliter  =  0.845  liquid  gill 

10  deciliters     =  1  liter  =  1.0567  liquid  quarts 

10  liters  =  1  dekaliter  —  2.6417  liquid  gallons 

10  dekaliters   =  1  hectoliter  =  2.8375  U.  S.  bushels 

10  hectoliters  =  1  kiloliter  (or  stere)  =  28.375  U.  S.  bushels 

Cubic  Measure 
1  cord  of  wood  =  a  pile  cut  4  feet  long,  piled  4  feet  high  and  8  feet  on  the 

ground  —  128  cubic  feet 
1  perch  of  stone  =  a  quantity  1  1/2  feet  thick,  1  foot  high  and  16  !/2  feet  long 

=  24  3/4  cubic  feet 

(NOTE. — A  perch  of  stone  is,  however,  often  computed  differently  in  different  locali- 
ties; thus,  in  most  if  not  all  of  the  States  and  Territories  west  of  the  Mississippi,  stone- 
masons figure  rubble  by  the  perch  of  16  1/2  cubic  feet.  In  Philadelphia,  22  cubic  feet  are 
called  a  perch.  In  Chicago,  stone  is  measured  by  the  cord  of  100  cubic  feet.  Check 
should  be  made  against  local  practice.) 

Board  Measure 

In  board  measure,  boards  are  assumed  to  be  one  inch  in  thickness.  Therefore,  feet 
board  measure  of  a  stick  of  square  timber  =  length  in  feet  X  breadth  in  feet  X  thickness 
in  inches. 


1-50  MATHEMATICS;   UNITS,   AND   SYMBOLS 

Shipping  Measure 

For  register  tonnage  or  measurement  of  the  entire  internal  capacity  of  a  vessel,  it  is 
arbitrarily  assumed,  to  facilitate  computation,  that: 

100  cubic  feet  =  1  register  ton 
For  the  measurement  of  cargo: 

40  cubic  feet  =  1  U.  S.  shipping  ton  =  32.143  U.  S.  bushels 

42  cubic  feet  =  1  British  shipping  ton  =  32.703  Imperial  bushels 

Dry  Measure 

One  IT.  S.  Winchester  bushel  contains  1.2445  cubic  feet  or  2150.42  cubic  inches.  It 
holds  77.601  pounds  distilled  water  at  62  deg  fahr. 

(NOTE. — The  above  is  a  struck  bushel.  A  heaped  bushel  in  general  equals  1  !/4  struck 
bushels,  although  for  apples  and  pears  it  contains  1.2731  struck  bushels  =  2737.72  cubic 
inches.) 

One  U.  S.  gallon  (dry  measure)  =  Vs  bushel  and  contains  268.8  cubic  inches. 

(NOTE. — This  is  not  a  legal  IT.  S.  dry  measure  and  therefore  is  given  for  comparison 
only.) 

One  British  Imperial  bushel  contains  1.2843  cubic  feet  or  2219.36  cubic  inches.  It  holds 
80  pounds  distilled  water  at  62  dog  fahr. 

One  British  Imperial  gallon  =  1/8  Imperial  bushel  and  contains  277.42  cubic  inches. 

1  Winchester  bushel  =  0.9694  Imperial  bushel 
1  Imperial  bushel       =*  1.032  Winchester  bushels 
Same  relations  as  above  maintain  for  gallons  (dry  measure) 

[NOTE.— 1  U.  S.  gallon  (dry)  =  1.164  U.  S.  gallons  (liquid).] 

U.  S.  Units 

2  pints         =  1  quart  =       67.2    cubic  inches 

4  quarts      =  1  gallon  *  =  8  pints  =    268.8    cubic  inches 

2  gallons  *  =  1  peck  =  16  pints  =  8  quarts  —     537.6    cubic  inches 

4  pecks        =  1  bushel  =  64  pints  =  32  quarts  =  8  gallons  *  =  2150.42  cubic  inches 
1  cubic  foot  contains  6.428  gallons  (dry  measure)  * 

Liquid  Measure 

One  U.  S-  gallon  (liquid  measure)  contains  231  cubic  inches.  It  holds  8.336  pounds 
distilled  water  at  62  deg  fahr. 

One  British  Imperial  gallon  contains  277.42  cubic  inches.  It  holds  10  pounds  distilled 
water  at  62  deg  fahr. 

1  U.  S.  gallon  (liquid)  =  0.8327  Imperial  gallon 

1  Imperial  gallon  =  1.201  U.  S.  gallons  (liquid) 

[NOTE.— 1  U.  S.  gallon  (liquid)  =  0.8594  U.  S.  gallon  (dry).] 

U.  S.  Units 

4  gills      =  1  pint  =     16  fluid  ounces 

2  pints     =  1  quart  =  8  gills  -     32  fluid  ounces 

4  quarts  =  1  gallon  =  32  gills  =  8  pints  =128  fluid  ounces 
1  cubic  foot  contains  7.4805  gallons  (liquid  measure) 

Apothecaries'  Fluid  Measure 

60  minims  =  1  fluid  drachm.  8  drachms  =  1  fluid  ounce 

In  the  TJ.  S.  a  fluid  ounce  is  the  128th  part  of  a  U.  S.  gallon,  or  1.805  cu  in.  or 
29.58  cu  cm.  It  contains  455.8  grains  of  water  at  62  deg  fahr.  In  Great  Britain  the  fluid 
ounce  is  1.732  cu  in.  and  contains  1  ounce  avoirdupois  (or  437.5  grains)  of  water  at  62  deg 
fahr. 

*  The  gallon  is  not  a  U.  S.  legal  dry  measure. 


Table  4.     Plane  Angle  [No  Dimensions] 


Multiply 

Number 

^N.         of-^ 

Revolu- 

t\>N^ 

Degrees 

Minutes 

Quadrants 

Radians  * 

(Circum- 

Seconds 

Obtain  ^sX* 

ferences) 

4-                   ^ 

Degrees 

1 

K667 

90 

57.30 

360 

2.778 

XI  0-2 

XlO-4 

Minutes 

60 

1 

5400 

3438 

2.16 

1.667 

xio* 

X10-2 

Quadrants 

Kill 

K852 

1 

0.6366 

4 

3.087 

X10-2 

XlO-4 

X10-6 

Radians  * 

K745 

2.909 

1.571 

1 

6.283 

4.848 

XI  0-2 

XlO-4 

XlO-c 

Revolutions  * 

2.778 

4.630 

0.25 

0,  1591 

1 

7.716 

(Circumferences) 

XIO-3 

X10-5 

X10-7 

Seconds 

3600 

60 

3.24X105 

2.063X105 

1.296X106 

1 

*  27T  radians  =  1  circumference  —  360  degrees  by  definition. 
Table  5.     Solid  Angle  [No  Dimensions] 


Multiply 
Number 
of-» 


Hemispheres 


Sphe: 


Spherical 
right  angles 


Steradians  t 


Hemispheres 


0.25 


0.  1592 


Spheres  * 


0.5 


0.125 


7.958  X  10-2 


Spherical  right  angles 


0.6366 


Steradians  f 


6.283 


12.57 


1.571 


*  A  sphere  is  the  total  solid  angle  about  a  point.       Lf  4ir  Steradians  =  1  sphere  by  definition. 

Table  6.     Time  [T] 


Multiply 
X  Number 
of—  ^ 
.^ 

Obtain         \X^ 

Days 

Hours 

Minutes 

Months 
(average)  * 

Seconds 

Weeks 

Days 

1 

4.167 
XI  0-2 

6.944 

30.42 

1.157 
XI  0-5 

7 

Hours 

24 

1 

1.667 
XI  0-2 

730.0 

2.778 

168 

Minutes 

1440 

60 

1 

4.380 

1.667 
X10-2 

K008 
X104 

Months  (average)  * 

3.288 
XI  0-2 

1.370 
XIO-3 

2.283 
XI  0-5 

1 

3.806 
X10-7 

0.2302 

Seconds 

8.64 
X104 

3600 

60 

2.628 
X106 

1 

6.048 
X105 

Weeks 

0.  1429 

5.952 
XI  0-3 

9.921 
XI  0-5 

4.344 

K654 
XI  0-6 

1 

*  One  common  year  =  365  days;  one  leap  year  =  366  days;  one  average  month  =  Ms  of  a 
common  year. 
1-51 

1-52  MATHEMATICS,   UNITS,   AND   SYMBOLS 

Table  7.     Linear  Velocity  [LT~l] 


v         Multiply 
v     >v       Number 

'"X^v 

Obtain     \N> 

4,        V\< 

S"d 

ii 

o 

Feet  per  minute 

1 

1 

i 

03     9 
|     fe 

1 

| 

Meters  per  second 

1 
1 

I 

Centimeters 
per  second 

1 

0.5080 

30.48 

27.78 

1667 

51.48 

1.667 

100 

44.70 

2682 

Feet  per  minute 

1.969 

1 

60 

54.68 

3281 

101.3 

3.281 

196.8 

88 

5280 

Feet  per  second 

3.281 

1.667 
XIO-2 

1 

0.9113 

54.68 

1.689 

5.468 
XIO-2 

3.281 

1.467 

88 

Kilometers  per  hour 

0.036 

1.829 
XIO-2 

1.097 

1 

60 

1.853 

0.06 

3.6 

1.609 

96.54 

Kilometers 
per  minute 

0.0006 

3.048 

xio-* 

1.829 
XIO-2 

1.667 
XIO-2 

1 

3.088 
XIO-2 

0.001 

0.06 

2.682 
XIO-2 

1.609 

Knots* 

1.943 
XIO-2 

9.868 
XI  0-3 

0.5921 

0.5396 

32.38 

1 

3.238 
XlO-a 

1.943 

0.8684 

52.10 

Meters  per  minute 

0.6 

0.3048 

18.29 

16.67 

1000 

30.88 

1 

60 

26.82 

1609 

Meters  per  second 

0.01 

5.080 
XI  0-3 

0.3048 

0.2778 

16.67 

0.5148 

1.667 

1 

0.4470 

26.82 

Miles  per  hour 

2.237 
XIO-2 

1.136 
XIO-2 

0.6818 

0.6214 

37.28 

1.152 

3.728 
XIO-2 

2.237 

1 

60 

Miles  per  minute 

3.728 
X10-4 

1.892 

1.136 
XIO-2 

1.036 
X10-a 

0.6214 

1.919 
XIO-2 

6.214 

3.728 
XIO-2 

1.667 
XIO-2 

1 

*  Nautical  miles  per  hour. 


The  Miner's  Inch 


(Used  in  Measuring  Flow  of  Water) 

An  Act  of  the  California  legislature,  May  23,  1901,  makes  the  standard  miner's  inch 
1.5  cu  ft  per  minute,  measured  through  any  aperture  or  orifice. 

The  term  miner's  inch  is  more  or  less  indefinite,  for  the  reason  that  California  water 
companies  do  not  all  use  the  same  head  above  the  center  of  the  aperture,  arid  the  inch 
varies  from  1.36  to  1.73  cu  ft  per  minute,  but  the  most  common  measurement  is  through 
an  aperture  2  in.  high  and  whatever  length  is  required,  and  through  a  plank  1  1/4  in.  thick. 
The  lower  edge  of  the  aperture  should  be  2  in.  above  the  bottom  of  the  measuring-box, 
and  the  plank  5  in.  high  above  the  aperture,  thus  making  a  6-in.  head  above  the  center 
of  the  stream.  Each  square  inch  of  this  opening  represents  a  miner's  inch,  which  is  equal 
.to  a  flow  of  1.5  cu  ft  per  minute. 


CONVERSION  TABLES 


1-53 


Table  8.     Angular  Velocity  [T*1] 


Multiply 
Number 
of-* 


Degrees 
per  second 


Radians 
per  second 


Revolutions 
per  minute 


Revolutions 
per  second 


Degrees  per  second 


57.30 


360 


Radians  per  second 


1.745X10-2 


0.1047 


6.283 


Revolutions  per  minute 


0.1667 


9.549 


60 


Revolutions  per  second 


2.778X10-3       0.1592 


.  667  X  IP"2 


Table  9.     Linear  Acceleration  *  [LT~2] 


Multiply 
-x.                 Number 
X\.        of-> 

t>^x. 
Obtain        ^xN^ 
•^                           >^ 

Centimeters 
per  second 
per  second 

Feet 
per  second 
per  second 

Kilometers 
per  hour 
per  second 

Meters 
per  second 
per  second 

Miles 
per  hour 
per  second 

Centimeters  per  second 
per  second 

1 

30.48 

27.78 

100 

44.70 

Feet  per  second  per 
second 

3.281  X  10-2 

1 

0.9113 

3.281 

1.467 

Kilometers  per  hour 
per  second 

0.036 

1.097 

i 

3.6 

1.609 

Meters  per  second  per 
second 

0.01 

0.3048 

0.2778 

1 

0.4770 

Miles  per  hour  per 
second 

2.237  X  10-2 

0.6818 

0.6214 

2.237 

, 

*  The  (standard)   acceleration  due  to  gravity   (#0)  =  980.7  cm  per  sec  per  sec   =  32.17  feet 
per  sec  per  sec  =  35.30  km  per  hour  per  sec  =  9.807  meters  per  sec  per  sec  =  21.94  miles  per  hour 

per  sec. 


Table  10.     Angular  Acceleration  [T~'2] 


Multiply 
Number 
of-> 


Radians 
per  second 
per  second 


Revolutions 
per  minute 
per  minute 


Revolutions 
per  minute 
per  second 


Revolutions 
per  second 
per  se§ond 


Radians  per  second  per  second 


.745X10-3 


0. 1047 


6.283 


Revolutions  per  minute  per  minute 


573.0 


1 


60 


3600 


Revolutions  per  minute  per  second 


9.549 


1.667X10-2 


60 


Revolutions  per  second  per  second 


0.1592 


2.778X10- 


1.667X10-2 


1-54 


MATHEMATICS,   UNITS,   AND   SYMBOLS 


Table  11.     Mass  [M]  and  Weight  * 


Multiply 
N.        Number 

to  ^\Vv 
Obtain      \^v 

1 

O 

1 

i 

! 

H— 

o 

j 

I 

1 

Tons  (short) 

Grains 

1 

15.43 

1.543 

xio4 

1.543 

437.5 

7000 

9.072 

105 

Grams 

6.481 
XIO'2 

1 

1000 

0.001 

28.35 

453.6 

1.016 

106 

Kilograms 

6.481 
XlO-s 

0.001 

1 

10-6 

2  835 
X10-2 

0  4536 

1016 

1000 

907.2 

Milligrams 

64.81 

1000 

106 

1 

2.835 

4.536 
X105 

1.016 
X109 

109 

9  072 
X1Q8 

Ounces  f 

2.286 

3.527 

35.27 

3.527 
X10-5 

1 

16 

3.584 

3.527 

3.2 

xio4 

Pounds  t 

1.429 

2.205 
XlO-3 

2.205 

2.205 
X10-6 

6.250 
XIO-2 

1 

2240 

2205 

2000 

Tons  (long) 

9.842 

9\842 
XI  0-4 

9.842 
XI  O-io 

2.790 

4.464 

1 

0.9842 

0.8929 

Tons  (metric) 

10^5 

0.001 

10-9 

2.835 
XI  0-5 

4  536 

1.016 

1 

0.9072 

Tons  (short) 

1.102 
X10-6 

-1.102 
XlO-3 

1  102 
XI  0"9 

3.125 
XI  0-s 

0.0005 

1.120 

1.102 

1 

*  These  same  conversion  factors  apply  to  the  gravitational  units  of  force  having  the  corresponding 
names.     The  dimensions  of  these  units  when  used  as  gravitational  units  of  force  are  MLT~*;  see 


table  for  Force. 

f  Avoirdupois  pounds  and  ounces. 

Metric  Multiples 
106   micrograms  =  103   milligrams  =  102    centigrams 


10"1  dekagram 
gram 


10  ~2  hectogram  =  10  3  kilogram  ; 


10    decigrams 
10"4  myriagram  =• 


=  1    gram  = 
10~8   mega- 


Avoirdupois  Weight 
(Used  Commercially) 
27.343  grains  =  1  drachm 

16  drachms  =  1  ounce  (oz)  =  437.5  grains 

16  ounces  =  1  pound  (Ib)  =  7000  grains 

28  pounds  =  1  quarter  (qr) 

4  quarters  —  1  hundredweight  (cwt)  =  112  pounds 

20  hundredweight  =  1  gross  or  long  ton  * 
2000  pounds  =  1  net  or  short  ton 

(*  NOTE. — The  long  ton  is  used  by  the  U.  S.  custom-houses  in  collecting  duties  upon 
foreign  goods.     It  is  also  used  in  freighting  coal  and  selling  it  wholesale.) 
14  pounds  =  1  stone;  100  pounds  =  1  quintal 

Troy  Weight 
(Used  in  weighing  gold  or  silver) 

24  grains  =  1  pennyweight  (dwt) 

20  pennyweights  =  1  ounce  (oz)  =  480  grains 
12  ounces  =  1  pound  (Ib)  =  5760  grains 

The  grain  is  the  same  in  Avoirdupois,  Troy  and  Apothecaries'  weights.     A  carat,  for 
weighing  diamonds  =  3.086  grains  =  0.200  gram.     (International  Standard,  19.13.) 
1  pound  troy  =    .8229  pound  avoirdupois 

1  pound  avoirdupois  =  1.2153  pounds  troy 


CONVEKSION  TABLES 


1-55 


Apothecaries'  Weight 
(Used  in  compounding  medicines) 
20  grains      —  1  scruple  O) 
3  scruples  =  1  drachm  (3 )  —  60  grains 
8  drachms  =  1  ounce  (§)  =  480  grains 
12  ounces     =  1  pound  (Ib)  =  5760  grains 

The  grain  is  the  same  in  Avoirdupois,  Troy  and  Apothecaries'  weights. 
1  pound  apothecaries  =  0.82286  pound  avoirdupois 
1  pound  avoirdupois    =  1.2153  pounds  apothecaries 

Table  12.     Density  or  Mass  per  Unit  Volume  [ML~3] 


Grams  per 

cubic 
centimeter 


Kilograms 

per 
cubic  meter 


Pounds  per 
cubic  foot 


Pounds  per 
cubic  inch 


Grams  per  cubic  centimeter 


0.001 


1  .602X  10-2 


27.68 


Kilograms  per  cubic  meter 


1000 


16.02 


2.768X104 


Pounds  per  cubic  foot 


62.43 


6.243X10-2 


1728 


Pounds  per  cubic  inch 


3.613X10-23.613X10-5 


5.787X10-4 


Pounds  per  mil  foot  * 


3.405  X  10~7  3.405X  10 -1°  5.456X  10-9  9.425X1Q-6 


*  Unit  of  volume  is  a  volume  one  foot  long  and  one  circular  mil  in  cross-section  area. 
Table  13.     Force  *  [MLT~2]  or  [F] 


Multiply 
s.            Number 

x\of-* 

Obtain   ^^"^^^ 

Dynes 

Grams 

Joules 
per  cm 

Joules 
per  meter 
(newtons) 

Kilo- 
grams 

Pounds 

Poundala 

Dynes 

1 

980.7 

107 

105 

9.807 
X105 

4.448 
X105 

1.383 
X104 

Grams 

1.020 
X\Q^S 

1 

1.020 
XI  04 

102.0 

1000 

453.6 

14.10 

Joules  per  cm 

10-7 

9.807 
XI  0-5 

1 

.01 

9.807 
XI  0-2 

4.448 
XI  0-2 

1.383 
X10-3 

Joules  per  meter 
(newtons) 

10-5 

9.807 
XlO-s 

100 

1 

9.807 

4.448 

0.  1383 

Kilograms 

1.020 
X\Q~* 

0.001 

10.20 

0.  1020 

1 

0.4536 

1.410 

Pounds 

2.248 

x\o-& 

2.205 
XI  0-3 

22.48 

0.2248 

2.205 

1 

3.  108 
X10-2 

Poundals 

7.233 
X10-5 

7.093 
X10-2 

723.3 

7.233 

70.93 

32.17 

1 

*  Conversion  factors  between  absolute 
acceleration  due  to  gravity  conditions. 


and  gravitational  units  apply  only  under  standard 


1-56 


MATHEMATICS,   UNITS,  AND   SYMBOLS 


Table  14.     Torque  or  Moment  of  Force  [MZ2T~2]  or  [FL]  * 


Multiply 
X.                   Number 

Obtain      ^^OsX 

Dyne- 
centimeters 

Gram- 
centimeters 

Kilogram- 
meters 

Pound-feet 

Newton- 
meter 

Dyne-centimeters 

1 

980.7 

9.  807  XIO7 

1.356  XIO7 

107 

Gram-centimeters 

1.020  XIO"3 

1 

105 

1.383  XIO4 

1.020  XIO4 

Kilogram-meters 

1.020  XIO"8 

10"6 

1 

0.1383 

0.1020 

Pound-feet 

7.  376  XIO"8 

7.233X10"5 

7,233 

1 

0.7376 

Newton-meter 

10"7 

9.807X10~4 

9.807 

1.305 

1 

*  Same  dimensions  as  energy. 


Table  15.     Pressure  or  Force  per  Unit  Area 


x       Multiply 

n-§ 

i 

>i 

-w     N.       Number 

* 

§  a 

§•43 

S  •«• 
^0 

1 

£ 

t&i 

£    JS 

pd 

«L 

.-§ 

^X\ 

1 

S  o 

11 

•So 

i0 

S1! 

Hi 

|.a 

5,  o 

f  J 

rd  <t> 

&8 

Obtain  ^VV 

4-        ^ 

1 

I  g 
&a 

m 

a  >» 

1  % 
§  ° 

P 

r 

i' 

•s  I 

I1 

P 

P 

II 
& 

Atmospheres  * 

1 

9.869 

xio-7 

1.316 
X10~2 

3.342 

xio~2 

2.458 

9.678 
X10"5 

4.725 
XIO"4 

6.804 
XIO"2 

0,9450 

9.869 
XIO"6 

Baryes  or  dynes  per 

•1.013 

1 

1.333 

3.386 

2.491 

98.07 

478.8 

6.895 

9.576 

10 

square  centimeter  f 

xio6 

xio4 

xio4 

XIO"3 

xio4 

X  105 

Centimeters  of  mer- 

76.00 

7.501 

1 

2.540 

0.1868 

7.356 

3.591 

5.171 

71.83 

7.501 

cury  at  0°  C  I 

XIO"5 

X10"3 

X10"2 

XIO"4 

Inches  of  mercury  at 

29.92 

2.953 

0.3937 

1 

7.355 

2.896 

1.414 

2.036 

28.28 

2.953 

o°ct 

xio~5 

XIO  2 

X10"3 

xio  2 

xio~4 

Inches   of  water  at 

406.8 

4.015 

5.354 

13.60 

1 

3.937 

0.1922 

27.68 

384.5 

4.015 

4°C 

X10"4 

XIO  2 

xio~3 

Kilograms  per  square 

1.033 

1.020 

136.0 

345.3 

25.40 

1 

4.882 

703.1 

9765 

0.  1020 

meter  § 

xio4 

xio  2 

Pounds  per  square 

2117 

2.089 

27.85 

70.73 

5.204 

0.2048 

1 

144 

2000 

2.089 

foot 

X10"3 

X  10"2 

Pounds  per  square 

14.70 

1.450 

0.1934 

0.4912 

3.613 

1.422 

6.944 

1 

13,89 

1,450 

inch     - 

X10"5 

xio~2 

X10~3 

XIO"3 

XIO"4 

Tons  (short)  per 

1.058 

1.044 

1.392 

3.536 

2.601 

1.024 

0.0005 

0.072 

1 

1.044 

square  foot 

X10"6 

X10"2 

xio~2 

X10"3 

X10"4 

XIO"6 

Newtoos  per  square 

1.013 

10"1 

1.333 

3.386 

2.491 

9.807 

47.88 

6.895 

9  576 

1 

meter 

xio5 

xio3 

xio3 

X10"4 

X  103 

XIO4 

*  Definition;  One  atmosphere  (standard)  =  76  cm  of  mercury  at  0  deg  cent, 
f  Sometimes  caBed  a  bar. 

t  To  convert  height  A  of  a  column  of  mercury  at  i  degrees  Centigrade  to  the  equivalent  height  A0  at  0  deg  cent  use 
^  =  h  { l  -  ™+J  }  where  m  =  0.0001818  and  I  =  18.4  X  10"6  if  the  scale  is  engraved  on  brass;  I  -  8.5  X  10~6 

luid)  see  International  Critical  Tables, 


§  1  gram  per  sq  em  =  10  kilograms  per  sq  m. 


CONVERSION   TABLES 


1-57 


Table  16.     Energy,  Work  and  Heat  *  [ML^T^]  or  [FL] 


Multiply 
N.         Number 

Obtain    ^"VV^ 

1 

S3 

1 

Ergs  or  centimeter- 
dynes 

£ 
1 

1 

S  1 

11 

j1"! 

j 

j 

I 

British  thermal 
units  t 

1 

9.297 
XI  0-8 

9.480 
XI  0-n 

1.285 
XI  0-3 

2545 

9.480 

3.969 

3413 

9.297 
XI  0-3 

3.413 

Centimeter-grams 

1.076 
X107 

1 

1.020 
XI  0-s 

1.383 
X1Q4 

2.737 
XI  Oio 

1.020 
X104 

4.269 
X107 

3.671 

XI  Oio 

105 

3.671 
X107 

Ergs  or  centimeter- 
dynes 

1.055 
XI  01° 

980.7 

1 

1.356 
X107 

2.684 
XI  013 

107 

4.186 
XI  Oio 

3.6 
XI  013 

9.807 
X107 

3.6 
XI  Oio 

Foot-pounds 

778.0 

7.233 
XI  0-6 

7.367 
X10-8 

1 

1.98 
X106 

0.7376 

3087 

2.655 
X106 

7.233 

2655 

Horsepower-hours 

3.929 
XI  0-4 

3.654 

x  10-11 

3.722 

5.050 
X10-7 

1 

3.722 
XI  0-7 

1.559 
XI  0-3 

1.341 

3.653 

1.341 
X10-3 

Joules  J  or  watt- 
seconds 

1054.8 

9.807 

10-7 

1.356 

2.684 
X1Q5 

1 

4186 

3.6 
X106 

9.807 

3600 

Kilogram-calories  t 

0.2520 

2.343 
X10-8 

2.389 

xi  0-11 

3.239 
XI  0-4 

641.3 

2.389 
X10-4 

1 

860.0 

2.343 
XI  0-3 

0.8600 

Kilowatt-hours 

2.930 
XI  0-4 

2.724 

xi  0-11 

2.778 
XI  0-14 

3.766 
XI  O-7 

0.7457 

2.778 
XI  0-7 

1.163 
XI  0-3 

1 

2.724 
'XI  O-e 

0.001 

Meter-kilograms 

107.6 

10-s 

1.020 
XI  0-8 

0.1383 

2.737 
X105 

0.1020 

426.9 

3.671 

I 

367.1 

Watt-hours 

0.2930 

2.724 
XlO-a 

2.778 

xi  0-11 

3.766 
XI  0-4 

745.7 

2.778 
X10-4 

1.163 

1000 

2.724 
XI  0-3 

1 

*  See  note  at  the  bottom  of  Table  17. 

t  Mean  calorie  and  Btu  used  throughout.  One  gram-calorie  =  0.001  kilogram-calorie;  one 
Oatwald  calorie  =  0.01  kilogram-calorie. 

The  IT  cal,  1000  international  steam-table  calories,  has  been  denned  as  the  1 /860th  part  of  the 
international  kilowatthour  (see  Mechanical  Engineering,  Nov.,  1935,  p.  710).  Its  value  is  very 
nearly  equal  to  the  mean  kilogram-calorie,  1  IT  cal  =  1.00037  ;kilogram-calories  (mean).  1  .Btu  =» 
251.996  IT  cal. 

t  Absolute  joule,  defined  as  107  ergs.  The  international  joule,  baaed  on  the  international  ohm. 
and  ampere,  equals  1,0003  absolute  joules, 


1-58  MATHEMATICS,   UNITS;   AND   SYMBOLS 


Multiply 
^    >v         Number 

to       \SsX 
Obtain    ^OO> 

British  thermal  units 
per  minute 

i 

1 

Foot-pounds  per 
second 

w 

W 

Kilogram-calories 
per  minute 

Kilowatts 

Metric  horsepower 

1 

British  thermal  units 
per  minute 

1 

5.689 
X10-9 

1.285 
X10-3 

7.712 
XIO-2 

42.41 

3.969 

56.89 

41.83 

5.689 
XIO-2 

Ergs  per  second 

1.758 

1 

2.259 
X1Q5 

1.356 
X107 

7.457 
X109 

6.977 
X108 

1010 

7.355 
X109 

107 

Foot-pounds  per 
minute 

778.0 

4.426 
X10-6 

1 

60 

3.3 

3087 

4.426 

xio* 

3.255 

44.26 

Foot-pounds  per 
second 

12.97 

7.376 
XI  0-8 

1.667 
XIO-2 

1 

550 

51.44 

737.6 

542.5 

0.7376 

Horsepower  * 

2.357 
XIO-2 

1.341 

3.030 
X10-5 

1.818 
X10-3 

1 

9.355 
XIO-2 

1.341 

0.9863 

1.341 
X10-3 

Kilogram-calories 
per  minute 

0.2520 

1.433 
X10-9 

3.239 

1.943 
XIO-2 

10.69 

1 

14.33 

10.54 

1.433 
XIO-2 

Kilowatts 

1.758 
XIO-2 

10-10 

2.260 
XI  0-5 

1.356 
XI  0-3 

0.7457 

6977 
XI  0-2 

1 

0.7355 

10-3 

Metric  horsepower 

2.390 
XIO-2 

1.360 

3.072 
XI  0-5 

1.843 
XI  0-3 

1.014 

9.485 
XI  0-2 

1.360 

1 

1.360 
XI  0-3 

Watts 

17.58 

10-7 

2.260 
XIO-2 

1.356 

745.7 

69.77 

1000 

735.5 

1 

1  Cheval-vapeur  =  75  kilogram-meters  per  second 
1  Poncelet  =  100  kilogram-meters  per  second 

*The  "horsepower"  used  in  these  tables  is  equal  to  550  foot-pounds  per  second  by  definition. 

Other  definitions  are  one  horsepower  equals  746  watts  (XI.  S.  and  Great  Britain)  and  one  horsepower 

equals  736  watts  (continental  Europe).     Neither  of  these  latter  definitions  is  equivalent  to  the  first; 

the  "horsepowers"  defined  in  these  latter  definitions  are  widely  used  in  the  rating  of  electrical 

machinery. 

Table  18.     Quantity  of  Electricity  and  Electric  Flux  [Q] 


Multiply 
V.                  Number 

Obtain      ^^Os>; 

Ab  coulombs 

Ampere- 
hours 

Coulombs 

Faradays 

Stat- 
coulomba 

Abcoulombs  * 

1 

360 

0.1 

9649 

3.335 

x  10-11 

Ampere-hours 

2.778 
XI  0-3 

1 

2.778 

xi  o-^ 

26.80 

9.259 
XI  0-" 

Coulombs 

10 

3600 

1 

9.649 

xio* 

3.335 

x  10-10 

Faradays 

1.036 
XI  0-* 

3.731 
XIO-2 

1.036 
XI  0-6 

1 

3.457 
X  10-16 

Statcoulombs  * 

2.998 
XI  Oio 

1.080 
XI  013 

2.998 
X109 

2,893 

1 

*  Conventionally  in  the  electrostatic  and  electromagnetic  systems  of  units  the  number  of  1inM  / 
electnc  flux  emanating  from  a  point  charge  is  4*  times  that  charge  (or  quantity  of  e^StrTcitvlTh  I 
statcoulomb  and  the  abcoulomb  are  units  of  charge  not  flux                    viuauwiy  01  electricity;,     ine 

CONVERSION  TABLES 


1-59 


Table  19.    Charge  per  Unit  Area  and  Electric  Flux  Density 


Multiply 
Number 

.x/^^x.                °f"^ 

Abcoulombs 

Coulombs 

Coulombs 

Statcoulombs 

Coulombs 

to^-^v. 

per  square 
centimeter  * 

per  square 
centimeter 

per  square 
inch 

per  square 
centimeter 

per  square 
meter 

Obtain     ^^^^^ 

Abcoulombs  per  square  centi- 
centimeter  * 

1 

0.1 

1.550X10~2 

3.335X1Q-11 

10~5 

Coulombs  per  square 
centimeter 

10 

1 

0.1550 

3.335X10"10 

io-4 

Coulombs  per  square  inch 

64.52 

6.452 

1 

2.151  X10~9 

6.452X10~4 

Statcoulombs  per  square 
centimeter  * 

2.998X1010 

2.998X109 

4.647X108 

1 

2.998  X105 

Coulombs  per  square  meter 

105 

10* 

1550 

3.335X10"6 

1 

*  See  footnote  to  Table  18. 


Table  20.    Electric  Current  [ 


Multiply 
Numbet 
of-» 


Abamperes 


Amperes 


Statamperes 


Abamperes 


0.1 


3.335  X  10~n 


Amperes 


10 


I 


3.335  X  10~10 


Statamperes 


2.998  X  1010 


2.998  X  109 


1 


Table  21.     Current  Density 


Multiply 
-x.                     Number 

^^v^^        of-* 

Abamperes 

Amperes 

Amperes 

Statamperes 

Amperes 

to^^X^ 

per  square 
centimeter 

per  square 
centimeter 

per  square 
inch 

per  square 
centimeter 

per  square 
meter 

Obtain       ^^\o^> 

Abamperes  per  square 

1 

0.1 

1.550X10~2 

3.335  X  10~u 

IO"6 

centimeter 

Amperes  per  square 
centimeter 

10 

1 

0.1550 

3.335X10~10 

,o-4 

Amperes  per  square  inch 

64.52 

6.452 

1 

2J51X10-*9 

6.452X10~4 

Statamperes  per  square 

2.998X1010 

2.998X109 

4.647X108 

1 

2.998X105 

centimeter 

Amperes  per  square  meter 

IO6 

IO4 

1550 

3.335X10~6 

1 

1-60  MATHEMATICS;   UNITS,   AND   SYMBOLS 

Table  22.    Electric  Potential  and  Electromotive  Force  {M Q-ltfT~z]  or  [FQ"1L] 


Multiply 
•s.                   Number 

Obtain       ^^^Ov^ 

Abvolts 

Microvolts 

Millivolts 

Statvolts 

Volts 

Abvolts 

1 

100 

105 

2.998 
XI  do 

108 

Microvolts 

0.01 

1 

1000 

2.998 
XI03 

108 

Millivolts 

10-5 

0.001 

1 

2.998 
X105 

1000 

Statvolts 

3.335 

3.335 

3.335 
XIO-6 

1 

3.335 

xio-» 

Volts 

10-8 

10-6 

0.001 

299.8 

1 

Table  23.     Electric  Field  Intensity  and  Potential  Gradient  [MQ-1LT~^]  or  [FQ~l] 


\  Multiply 

\    \Number 

Abvolts 
per 
centi- 
meter 

Micro- 
volts 
per 
meter 

Milli- 
volts 
per 
meter 

Statvolts 
per 
centi- 
meter 

Volts 
per 
centi- 
meter 

Kilo- 
volts 
per 
centi- 

Volts 
per 
inch 

Volts 
per 
mil 

Volts 
per 
meter 

Obtain   \V 

meter 

4>      ^ 

Abvolts  per 

1 

1 

1000 

2.998 

108 

10u 

3.937 

3.937 

10° 

centimeter 

XIO10 

xio7 

X  1010 

Microvolts  per 

I 

1 

1000 

2.998 

108 

10U 

3.937 

3.937 

106 

meter 

XlQi° 

xio7 

X  1010 

Millivolts  per 

0.001 

0.001 

1 

2.998 

105 

10s 

3.937 

3.937 

1000 

meter 

xio7 

X  104 

xio7 

Statvolts  per 

3.335 

3.335 

3.335 

1 

3.335 

3.335 

1.313 

1.313 

3.335 

centimeter 

x  10 

x  io~u 

X10~8 

X10~3 

X  I0~5 

Volts  per 
centimeter 

10- 

10- 

10~5 

299.8 

1 

1000 

0.3937 

393.7 

10~2 

Klovolts  per 
centimeter 

10-11 

10-11 

10-8 

0.2998 

0.001 

1 

3.937 
X  I  Or* 

0.3937 

JQ-5 

Volts  per  inch 

2.540 
XIO"8 

2.540 
X  10"8 

2.540 
X  10~5 

761.6 

2.540 

2540 

1 

1000 

2.540 
X  10"2 

Volts  per  mil 

2.540 

x  10-11 

2.540 
X  10~11 

2.540 

0.7616 

2.540 

xio-3 

2.540 

0.001 

1 

2.540 
X  10~5 

Volts  per  meter 

10"6 

10-6 

10   3 

2.998 

100 

105 

39.37 

3.937 

j 

XIO4 

xio4 

CONVERSION   TABLES 
Table  24.     Electric  Resistance  [MQ~2L2T~ll  or  [FQ~*LT\ 


1-61 


Multiply 
,.                      Number 

Obtain      ^^^$^1 

Abohma 

Megohms 

Microhms 

Ohms 

Statohms 

Abohms 

\ 

1015 

1000 

109 

8.988 
X1020 

Megohms 

10-15 

1 

10-12 

10-6 

8.988 
X105 

Microhms 

0.001 

1012 

1 

106 

8.988 
XI  017 

Ohms 

10-9 

106 

10-6 

1 

8.988 

xi  on 

Statohms 

1.112 
XI  0-21 

1.112 
X10-6 

1.112 
XI  0-18 

1.112 
XI  0-" 

I 

Electrical  Conductance  [F^1QL 
1  mho  =  1  ohm"1  ==  10~6  megmho  =  10s  micromho 


Table  25.     Electric  Resistivity  *    [MQ~2LZT~-1]  or  [FQ~*L*T] 


Multiply 
V.                  Number 

Obtain      ^^O^, 

Abohm- 
centimeters 

Microhm- 
centi- 
meters 

Microhm- 
inches 

Ohms 
(mil,  foot) 

Ohms 
(meter, 
gram)  t 

Ohm- 
meters 

Abohm-centimeters 

' 

1000 

2540 

166.2 

s 

,0u 

Microhm-centimeters 

0.001 

1 

2.540 

0.1662 

]00 
8 

108 

Microhm-inches 

3.937X  10~4 

0.3937 

1 

6.545X  10~2 

39.37 
5 

3.937X  107 

Ohms  (mil,  foot) 

6.015  X  10~3 

6.015 

15.28 

1 

601.5 
S 

6.015  X  108 

Ohms  (meter,  gram)  f 

10-« 

0.015 

2.540 
X  10-25 

1.662 
X  10~35 

1 

10-, 

Ohm-meters 

io-n 

10-8 

2.540X  10~8 

1.662X10"9 

10-6 

5 

1 

*  In  this  table  S  is  density  in  grams  per  cm.3  The  following  names,  corresponding  respectively  to  those  at  the  tops 
of  columns,  are  sometimes  used:  abohms  per  cm  cube;  microhms  per  cm  cube;  microhms  per  inch  cube;  ohms  per  mil- 
foot;  ohms  per  meter-gram.  The  first  four  columns  are  headed  by  units  of  volume  resistivity,  the  last  by  a  unit  of  mass 
resistivity.  The  dimensions  of  the  latter  are  Q~*L6T~l;  not  these  given  in  the  heading  of  the  table. 

t  One  ohm  (meter,  gram)  =  5710  ohms  (mile,  pound). 


1-62 


MATHEMATICS,   UNITS,  AND   SYMBOLS 


Table  26.     Electric  Conductivity*     [M^Q^L^T]  or  [F~1Q2L~*T 


Multiply 

.                       Number 

>^\1 

Abmhos 
per  cm 

Mhos 
(mil,  foot) 

Mhos 
(meter, 
gram) 

Micro- 
mhos 
per  cm 

Micro- 
mhos 
per  inch 

Mhos 
per 

meter 

Obtain       r^^^sj 

Abmhos  per  cm 

1 

6.015 

10~55 

0.001 

3.937 

,0~n 

X  10~3 

X  10   4 

Mhos  (mil,  foot) 

166.2 

1 

1.662 

0.1662 

6.524 

1.662 

X  10~35 

X  10~2 

x  io~9 

Mhos  (meter,  gram) 

105/5 

60  1  .  5/5 

1 

100/5 

39.37/5 

10-  '/« 

Micromhos  per  cm 

1000 

6.015 

0.015 

1 

0.3937 

I0~8 

Micromhos  per  inch 

2540 

15.28 

2.540 

2.540 

1 

2.54 

X  10~25 

X  10~8 

Mhos  per  meter 

1011 

6.015 

1065 

108 

3.937 

1 

X  108 

X  107 

*  See  footnote  of  Table  25,  Electric  Resistivity.    Names  sometimes  used  are  abmho  per  cm  cube, 
mho  per  mil-foot,  etc.    Dimensions  of  mass  conductivity  are  Q2IT"6T. 


Table  27.     Capacitance  [M-iQ*L~2T2]  or  [ 


Abfarada 


Farads 


Microfarads 


Statfarada 


Abfarada 


10-9 


10-15 


1.112 

X  10-21 


Farads 


109 


10-e 


1.112 
X  10-12 


Microfarads 


Statfarads 


1015 


106 


8.988 
X  1020 


8.988 


8.988 
X  105 


1.112 
X  10-fi 


CONVERSION  TABLES 
Table  28.     Inductance  [MQ~ZLZ]  or  [FQ~*LT*] 


1-63 


Multiply 


Abhenrys  * 


IO9 


1000 


IO6 


8.988 
X  IO20 


Henrys 


io 


10~6 


0.001 


8.988 
X  IO11 


Microhenrys 


0.001 


IO6 


1000 


8.988 
X  IO17 


Millihenrys 


10~6 


1000 


0.001 


8.988 
X  IO14 


Stathenrys 


1.112 
X  10~2> 


1.112 
X  10~12 


1.112 
X  10~18 


1.112 
X  10~15 


*  An  abhenry  is  sometimes  called  a  "centimeter."     See  footnote  to  Table  30  on  "Magnetic  Flux 
Density." 


Table  29.     Magnetic  Flux 


Multiply 
Number 


Kilolines 


Maxwells 
(or  lines) 


Webers 


Kilolines 


0.001 


106 


Maxwells  (or  lines) 


1000 


Webers 


10-s 


10-8 


Table  30.     Magnetic  Flux  Density  [MQ^T^]  or 


Multiply 

-x^^^                 Number 

Obtain       ^^O^L. 

Gausses 
(or  lines 
per  square 
centimeter) 

Lines  per 
square  inch 

Webers 
per  square 
centimeter 

Webers 
per  square 
inch 

Webers 
per  square 
meter 

Gausses  (or  lines  per 

1 

0.1550 

IO8 

1.550 

IO4 

square  centimeter) 

X  IO7 

Lines  per  square  inch 

6.452 

1 

6.452 

IO8 

6.452 

X  10s 

X  IO4 

Webers  per  square 

i<r8 

1.550 

1 

0.1550 

10-4 

centimeter 

x  io~9 

Webers  per  square  inch 

6.452 

10~8 

6.452 

1 

6.452 

X  10~8 

X  10~4 

Webers  per  square  meter 

io-4 

1.550 

IO4 

1550 

1 

X  10~6 

1-64  MATHEMATICS,   UNITS,   AND  SYMBOLS 

Table  31.    Magnetic  Potential  and  Magnetomotive  Force  [Qf1] 


Multiply 
Number 


Abampere-turns 


Ampere-turns 


Gilberts 


Abampere-turns 


0.1 


7.958  X  10-2 


Ampere-turns 


10 


0.7958 


Gilberts 


12.57 


1.257 


Table  32.    Magnetic  Field  Intensity,  Potential  Gradient,  and  Magnetizing  Force  [QL  1T  l] 


Multiply 
Number 


Abampere- 
turns  per 
centimeter 


Ampere- 
turns  per 
centi- 
meter 


Ampere- 
turns  per 
inch 


Oersteds 
(gilberts 
per  centi- 
meter) 


Ampere- 
turns  per 
meter 


Abampere-turns  per  centimeter 


0.1 


3.937 
X  10~2 


7.958 
X  10~2 


Ampere-turns  per  centimeter 


0.3937 


0.7958 


Ampere-turns  per  inch 


25.40 


2.540 


2.021 


2.54 
X  10~2 


Oersteds  (gilberts  per  centimeter) 


12.57 


1.257 


0.4950 


1.257 
X  10~2 


Ampere-turns  per  meter 


103 


102 


39.37 


79.58 


Table  33.    Specific  Heat 

(t  =  temperature) 

To  convert  specific  heat  in  any  unit  given  to  any  other  unit  multiply  the  number  of  original  units 
by  a  factor  obtained  by  dividing  the  factor  in  the  last  column  for  the  final  unit  by  the  factor  for  the 
original  unit. 


Unit  of  Heat  or  Energy 

Unit  of  Mass 

Temperature  Scale* 

Factor 

Gram-calories  

Gram 
Kilogram. 
Pound 
Pound 
Gram 
Pound 
Kilogram 
Pound 

Centigrade 
Centigrade 
Centigrade 
Fahrenheit 
Centigrade 
Fahrenheit 
Centigrade 
Fahrenheit 

1 
1 
1.800 
1.000 
4.186 
1055. 
1.163  X  10-8 
2.930  X  10~* 

Kilngrrftm-fifl.lnripR.  T  ,.,,,.,..  ,  .  .  ,  .  . 

British  thermal  units  

Joules  ,  

Kilowatt-hours  

Kilowatt-hours  

*  Temperature  conversion  formulas: 

tc  s=  temperature  in  Centigrade  degrees 
tf  =  temperature  in  Fahrenheit  degrees 
1  deg  fahr  =  (§6)  deg  cent. 
-  32) 


CONVERSION    TABLES 


1-65 


Table  34.     Thermal  Conductivity  [MLT~^-1]  and  Thermal  Resistivity  [M-lL~lT*t] 

(t  =  temperature) 

To  convert  thermal  conductivity,  in  gram-calories  transmitted  per  second  from  one  face  of  a  cube 
1  cm  on  edge  to  the  opposite  face  per  degree  centigrade  temperature  difference  between  these  faces, 
to  the  units  given  in  any  line  of  the  following  table,  multiply  by  the  factor  in  the  last  column. 

To  convert  thermal  conductivity  in  any  unit  given  to  any  other  unit  multiply  the  number  of  original 
units  by  a  factor  obtained  by  dividing  the  factor  in  the  last  column  for  the  final  unit  by  the  factor  for 
the  original  unit. 

To  convert  thermal  resistivity,  in  degrees  centigrade  between  one.  face  of  a  cube  1  cm  on  edge  and 
the  opposite  face  per  gram-calories  transmitted  per  second  between  these  faces,  to  the  units  given  in 
any  line  of  the  following  table,  divide  by  the  factor  in  the  last  column. 

To  convert  thermal  resistivity  in  any  given  unit  to  any  other  unit  multiply  the  number  of  the  original 
units  by  a  factor  obtained  by  dividing  the  factor  in  the  last  column  for  the  original  unit  by  the  factor 
for  the  final  unit. 

Surface  emission  resistance  in  thermal  ohms  per  square  centimeter  is  derived  from  degrees  f ahrenheit 
per  Btu  per  hour  per  square  foot  by  multiplying  the  number  of  the  latter  units  by  1761. 


Units  of 

Temperature 
Scale 

Factor 

Heat 

Area 

Thickness 

Time 

Gram-calories 

cm2 
m2 
ft2 
cm2 
ft2 
m2 
ft2 

cm 
cm 
inch 
cm 
inch 
cm 
inch 

second 
hour 
hour 
second 
second 
hour 
hour 

Centigrade 
Centigrade 
Fahrenheit 
Centigrade 
Fahrenheit 
Centigrade 
Fahrenheit 

1 
3.6  X  104 
2903. 
4.186 
850.6 
41.86 
0.8506 

TCi  1  ogrftm  -Oftl  oripR 

Joules  *       ...                      

Kilowatt-hours                .        .... 

Kilowatt-hours  

*  Thermal  resistances  in  these  units  are  known  as  thermal  ohms. 


Table  35.     Light 


Multiply 

^        Number 

^\T  of-^ 

Inter- 
national 
candles 

Hefners 

10-cp 
pentanes 

Carcels 

Bougie 
deci- 
males 

English 
candles 

German 
candles 

Obtain      ^^\J 

4-                         >± 

International  candles 

1.00 

0.90 

10.0 

9.61 

1.00 

1.04 

1.055 

Hefners 

1.11 

1.00 

11.1 

10.66 

1.11 

1.154 

1.17 

10-cp  pentanes 

0.10 

0.09 

1.00 

0.96 

0.10 

0.104 

0.105 

Carcels 

0.104 

0.094 

1.04 

1.00 

0.  104 

0.1 

0.109 

Bougie  decimales 

1.00 

0.90 

10.0 

9.61 

1.00 

1.04 

1.055 

English  candles 

0.96 

0.864 

9.6 

9.24 

0.96 

1.00 

1.02 

German  candles 

0.95 

0.855 

9.5 

9.19 

0.95 

0.98 

1.00 

1-66  MATHEMATICS,  UNITS,  AND  SYMBOLS 

18.  GAGES 

SHEET  METAL  GAGES.  The  important  sheet  metal  gages  in  use  in  the  United  States 
are-  the  United  States  Standard  Gage  for  sheet  and  plate  iron  and  steel,  the  American 
Wire  Gage  (also  called  the  Brown  and  Sharpe  W.G.)  for  copper,  aluminum,  and  brass  and 
other  non-ferrous  alloys,  the  Tin  Plate  Gage,  the  Galvanized  Sheet  Gage,  the  American 
Zinc  Gage,  and  the  Birmingham  Wire  (or  Stubs'  Iron  Wire)  Gage.  In  Canada  and  Eng- 
land the  Birmingham  Gage  (different  from  the  Birmingham  Wire  Gage)  and  tho  Imperial 
Standard  Wire  Gage  (S.W.G.)  are  used.  Still  other  gages  are  used  elsewhere.  In  Japan 
standard  thickness  of  sheet  metal  is  denoted  by  the  thickness  in  millimeters.  A  standard 
Decimal  Gage,  in  which  the  standard  thicknesses  are  denoted  by  decimal  parts  of  an 
inch  and  not^by  gage  numbers,  has  been  used  in  the  United  States.  Copper  sheets  may 
be  obtained  with  thicknesses  any  integral  multiple  of  Vie  of  an  inch  up  to  2  in.  Heavy 
copper  sheets  may  be  obtained  in  definite  weights  per  square  foot.  Each  ounce  of  weight 
is  equivalent  to  approximately  0.001352  in.  thickness.  Lead  is  usually  ordered  in  this 
manner,  each  pound  being  equivalent  to  approximately  0.017  in.  thickness. 

The  United  States  Standard  Gage  for  sheet  iron  and  steel  (Act  of  Congress,  March  3, 
1893;  formerly  the  legal  standard  for  duties)  is  a  weight  gage  based  on  a  density  for  wrought 
iron  of  480  pounds  per  cubic  foot.  Since  1893,  steel  (density  of  489.6  Ib  per  cu  ft)  has 
come  into  general  use.  A  given  gage  number  of  this  gage  represents  a  fixed  weight  per 
unit  area;  hence  a  steel  sheet  will  have  a  smaller  thickness  than  a  wrought  iron  sheet  of 
the  same  gage  number.  Monel  metal  sheets  are  rolled  to  the  thickness  given  for  wrought 
iron  without  regard  to  its  weight,  which  is  about  552.2  Ib  per  cu  ft.  Practice  among  steel 
manufacturers  is  irregular,  some  keeping  the  thickness  constant  for  a  given  gage  number 
irrespective  of  weight.  If  this  practice  is  followed,  the  weight  per  square  foot  and  per 
square  meter  given  in  the  second  and  third  columns  of  Table  36  will  vary,  whereas  thick- 
ness will  remain  near  that  given  for  wrought  iron. 

The  American  Wire  Gage  specifies  thicknesses  without  regard  to  weight.  For  the 
basis  of  this  gage  see  Wire  Gages,  p.  1-70,  where  are  also  given  the  Birmingham  W.G 
and  the  S.W.G. 

Tables  of  Thickness  and  Weight  corresponding  to  United  States  Standard  gage  and 
American  Wire  gage  numbers  are  shown  in  Tables  36  and  37.  These  tables  are  taken 
from  Circular  391  of  the  Bureau  of  Standards,  in  which  are  given  all  the  gages  mentioned 
above  and  the  tolerances  customary  in  commerce. 

WIRE  GAGES.  The  sizes  of  wires  having  a  diameter  less  than  1/2  in.  are  usually  stated 
in  terms  of  certain  arbitrary  scales  called  "gages."  The  size  or  gage  number  of  a  solid 
wire  refers  to  the  cross-section  of  the  wire  perpendicular  to  its  length;  the  size  or  gage 
number  of  a  stranded  wire  refers  to  the  total  cross-section  of  the  constituent  wires,  irre- 
spective of  the  pitch  of  the  spiraling.  Larger  wires  are  usually  described  in  terms  of  their 
area  expressed  in  circular  mils.  A  circular  mil  is  the  area  of  a  circle  1  mil  in  diameter,  and 
the  area  of  any  circle  in  circular  mils  is  equal  to  the  square  of  its  diameter  in  mils. 

There  are  a  number  of  wire  gages  in  use,  the  principal  ones  being  the  following: 

American  or  Brown  and  Sharpe  Wire  Gage.  This  gage  is  the  one  commonly  used  in 
the  United  States  for  copper,  aluminum,  and  resistance  wires.  The  gage  is  designated  by 
either  of  the  abbreviations  A.W.G.  or  B.  &  S. 

Basis  of  the  A.W.G.  or  B.  &  S.  Gage.  The  diameters  of  wires  having  successive  num- 
bers on  this  gage  are  in  the  ratio  of  V92(  «  1.1229  approx.)  to  1,  and  the  No.  36  wire  has 
a  diameter  of  5  mils.  No.  35  A.W.G.,  therefore,  has  a  diameter  of  5  X  1.1229  «  5.61  mils, 
and  so  on  until  No.  0000  is  reached,  having  a  diameter  of  460  mils. 

The  ratio  V^92  is  approximately  equal  to  V^,  which  is  1.1225.  This  circumstance 
makes  it  possible  to  have  a  group  of  wires  of  regular  gage  size  with  an  aggregate  area 
approximately  equal  to  that  of  another  regular  gage  size.  For  example,  a  reduction  of 
three  gage  numbers  (as  from  gage  No.  36  to  No.  33)  results  in  a  new  gage  number  repre- 
senting a  diameter  approximately  V%  times  that  represented  by  the  original  gage  num- 
ber— or  an  area  approximately  two  times  as  great. 

The  following  approximate  relations  are  also  useful: 

An  increase  of  1  in  the  number  increases  the  resistance  25  per  cent. 
An  increase  of  2  in  the  number  increases  the  resistance  60  per  cent 
An  increase  of  3  in  the  number  increases  the  resistance  100  per  cent. 
An  increase  of  10  in  the  number  increases  the  resistance  10  times 


GAGES 


1-67 


Table  36.   United  States  Standard  Gage  *  for  Sheet  and  Plate  Iron  and  Steel,  and  Its 

Extension  f 


Gage  No. 

Weight 
per  square  foot 

Weight 
per 
square 
meter 

Approximate  thickness 

Wrought  iron 
480  Ib/ft* 

Steel  and  open- 
hearth  iron 
489.6  Ib/ft* 

Ounces 

Pounds 

kg 

Inch 

mm 

Inch 

mm 

0000000.... 
000000..    .. 

320 
300 
280 
260 
240 

220 
200 
180 
170 
160 

150 
140 
130 
120 
110 

100 
90 
80 
70 
60 

50 
45 
40 
36 
32 

28 
24 
22 
20 
18 

16 
14 
12 
11 
10 

9 
8 
7 

6V2 
6 
51/2 
5 
41/2 

41/4 
4 
33/4 
31/2 
33/8 
31/4 
31/8 
3 

20.00 
18.75 
17.50 
16.25 
15.00 

13.75 
12.50 
11.25 
10.62 
10.00 

9.375 
8.750 
8.125 
7.500 
6.875 

6.250 
5.625 
5.000 
4.375 
3.750 

3.125 
2.812 
2.500 
2.250 
2.000 

1.750 
1.500 
1.375 
1.250 
1.125 

1.000 
.8750 
.7500 
.6875 
.6250 

.5625 
.5000 
.4375 
.4062 

.3750 
.3438 
.3125 
.2812 

.2656 
.2500 
.2344 
.2188 

.2109 
.2031 
.1953 
.1875 

97.65 
91.55 
85.44 
79.34 
73.24 

67.13 
61.03 
54.93 
51.88 
48.82 

45.77 
42.72 
39.67 
36.62 
33.57 

30.52 
27.46 
24.41 
21.36 
18.31 

15.26 
13.73 
12.21 
10.99 
9.765 

8.544 
7.324 
6.713 
6.103 
5.493 

4.882 
4.272 
3.662 
3.357 
3.052 

2.746 
2.441 
2.136 
1.983 

.831 
.678 
.526 
.373 

.297 
.221 
.144 
.068 

1.030 
.9917 
.9536 
.9155 

0.500 
.469 
.438 
.406 
.375 

.344 
.312 
.2812 
.2656 
.2500 

.2344 
.2188 
.2031 
.1875 
.1719 

.1562 
.1406 
.1250 
.1094 
.0938 

.0781 
.0703 
.0625 
.0562 
.0500 

.0438 
.0375 
.0344 
.0312 
.0281 

.0250 
.0219 
.0188 
.0172 
.0156 

.0141 
.0125 
.0109 
.0102 

.0094 
.0086 
.0078 
.0070 

.0066 
.0062 
.0059 
,0055 

.0053 
.0051 
.0049 
.0047 

12.70 
11.91 
11.11 
10.32 
9.52 

8.73 
7.94 
7.14 
6.75 
6.35 

5.95 
5.56 
5.16 
4.76 
4.37 

3,97 
3.57 
3.18 
2.778 
2.381 

.984 
.786 
.588 
.429 
.270 

1.111 
.952 
.873 
.794 
.714 

.635 
.556 
.476 
.437 
.397 

.357 
.318 
.278 
.258 

.238 
.218 
.198 
.179 

.169 
.159 
.149 
.139 

.134 
.129 
.124 
.119 

0.490 
.460 
.429 
.398 
.368 

.337 
.306 
.2757 
.2604 
.2451 

.2298 
.2145 
.1991 
.1838 
.1685 

.1532 
.1379 
.1225 
.1072 
.0919 

,0766 
.0689 
.0613 
.0551 
.0490 

.0429 
.0368 
.0337 
.0306 
.0276 

.0245 
.0214 
.0184 
.0169 
.0153 

.0138 
.0123 
.0107 
.0100 

.0092 
.0084 
.0077 
.0069 

.0065 
.0061 
.0057 
.0054 

.0052 
.0050 
.0048 
.0046 

12.45 
11.67 
10.90 
10.12 
9.34 

8.56 
7.78 
7.00 
6.62 
6.23 

5.84 
5.45 
5.06 
4.67 
4.28 

3.89 
3.50 
3.11 
2.724 
2.335 

1.946 
1.751 
1.557 
1.400 
1.245 

1.090 
.934 
.856 
.778 
.700 

.623 
.545 
.467 
.428 
.389 

.350 
.311 
.272 
.253 

.233 

.214    , 
.195 
.175 

.165 
.156 
.146 
.136 

.131 
.126 
.122 
.117 

00000  

oooo    

000  

00  . 

o  

1  

2  

3  

4  

5  

6  

7  

8  

9  

10  . 

11  

12 

13  

14  

15  

16  

17 

18  

19  

20  

21   

22  

23  

24  

25.. 

26  

27  

28  

29  .    ...  

30  

31    

32  

33  

34  

35  

36  

37  

38  

39  

40  

41  

42  

43... 

44  

*  For  the  Galvanized  Sheet  Gage,  add  2.5  ounces  to  the  weight  per  square  foot  as  given  in  the 
table.    Gage  numbers  below  8  and  above  34  are  not  used  in  the  Galvanized  Sheet  Gage. 

t  Gage  numbers  greater  than  38  were  not  in  the  standard  as  set  up  by  law,  but  are  in  general  use. 


1-68 


MATHEMATICS,   UNITS,   AND   SYMBOLS 


Table  37.    American  Wire  Gage — Weights  of  Copper,  Aluminum,  and  Brass  Sheets  and 

Plates 


Gage  No. 

Thickness 

Approximate  weight  *  per  sq  ft  in  Ib 

Inch 

JDTYl 

Copper 

Aluminum 

Commercial 
(high)  brass 

0000           

0.4600 
.4096 
.3648 
.3249 
.2893 

.2576 
.2294 
.2043 
.1819 
.1620 

.1443 
.1285 
.1144 
.1019 

,0907 
.0808 
.0720 
.0641 

.0571 
.0508 
.0453 
.0403 

.0359 
.0320 
.0285 
.0253 
.0226 

.0201 
.0179 
.0159 
.0142 
.0126 

,0113 
.0100 
.00893 
.00795 

.00708 
.00630 
.00561 
.00500 

.00445 
.00397 
.00353 
.00314 

11.68 
10.40 
9.266 
8.252 
7.348 

6.544 
5.827 
5.  189 
4.621 
4.115 

3.665 
3.264 
2.906 
2.588 

2.305 
2.053 
1.828 
1.628 

1.450 
1.291 
1.150 
1.024 

0.9116 
,8118 
.7230 
.6438 
.5733 

.5106 
.4547 
.4049 
.3606 
.3211 

.2859 
.2546 
.2268 
.2019 

.1798 
.1601 
.1426 
,1270 

.1131 
.1007 
.0897 
.0799 

21.27 
18.94 
16.87 
15.03 
13.38 

11.91 
10.61 
9.45 
8.41 
7.49 

6.67 
5.94 
5.29 
4.713 

4.195 
3.737 
3.330 
2.965 

2.641 
2.349 
2.095 
1.864 

.660 
.480 
.318 
.170 
.045 

0.930 

.828 
.735 
.657 
.583 

.523 
.4625 
.4130 
.3677 

.3274 
.2914 
.2595 
.2312 

.2058 
.1836 
.1633 
.1452 

6.49 
5.78 
5.  14 
4.58 
4.08 

3.632 
3.234 
2.880 
2.565 
2.  284 

2.034 
1.812 
.613 
.437 

.279 

.139 
.015 
0.904 

.805 
.716 
.639 
.568 

.506 
.451 
.402 
.3567 
.3186 

.2834 
.2524 
.2242 
.2002 
.1776 

.1593 
.1410 
.1259 
.1121 

.0998 
.0888 
.0791 
.0705 

.0627 
.0560 
.0498 
.0443 

20.27 
18.05 
16.07 
14.32 
12.75 

11.35 
10.11 
9.00 
8.01 
7.14 

6.36 
5.66 
5.04 
4.490 

3.996 
3.560 
3.  172 
2.824 

2.516 
2.238 
.996 
.776 

.582 
.410 
.256 
.115 
0.996 

.886 
.789 
.701 
.626 
.555 

.498 
.4406 
.3935 
.3503 

.3119 
.2776 
.2472 
.2203 

.1961 
.1749 
.1555 
.  1383 

000 

00             

0                         ... 

1  

2       

3 

4                     

5                       ... 

6  

7 

8   

9 

10   

11  

12             

13 

14  

15 

16   .. 

17 

18  

19  

20 

21 

22   .. 

23 

24   . 

25 

26   ..                .      . 

27 

28  

29 

30  

31  

32  

33   .          .      ..... 

34 

35... 

36 

37  

38  

39  

40  

Cubi°  centimeter' 


•,  8.89;  aluminum. 


GAGES  1-69 

A  No.  10  A.W.G.  copper  wire  has  the  following  approximate  characteristics: 

Ohms  per  1000  ft 1 

Circular  mils  area 10,000 

Weight,  pounds  per  1000  ft 32 

A  No.  10  A.W.G.  aluminum  wire  has  the  following  approximate  characteristics: 

Ohms  per  1000  ft 1.6 

Circular  mils  area 10,000 

Weight,  pounds  per  1000  ft 9.5 

Remembering  these  rules  it  is  easy  to  find  the  approximate  size,  resistance,  area,  or 
weight  of  any  size  wire.  For  example,  a  No.  12  A.W.G.  copper  wire  has  a  resistance  of 
1  plus  60  per  cent  =  1.6  ohms  per  1000  ft  approximately.  Its  area,  being  inversely  as 
its  resistance,  is  10,000/1.6  =  6250  circular  mils;  its  diameter  is  therefore  \/6250  =  79 
mils,  and  its  weight  32/1.6  =  20  Ib  per  1000  ft. 

U.  S.  Steel  Wire  Gage.  This  gage,  known  also  as  the  "Washburn  and  Moen,"  "Roeb- 
ling,"  "American  Steel  and  Wire  Co.'s  gage,"  is  the  one  usually  employed  in  the  United 
States  for  steel  and  iron  wire.  It  is  frequently  abbreviated  "S.W.G.,"  but  to  avoid  con- 
fusion with  the  British  Standard  Wire  Gage  (see  below]  it  should  be  abbreviated  "StL 
W.G."  or  "A.  (steel)  W.G." 

Birmingham  (or  Stubs'  Iron)  Wire  Gage.  This  gage  is  still  used  in  the  United  States 
for  some  purposes,  e.g.,  to  designate  the  size  of  brass  wire,  and  is  also  employed  to  a 
limited  extent  in  Great  Britain.  It  is  usually  abbreviated  "B.W.G."  It  is  sometimes 
referred  to  as  the  "Stubs'  Iron  Wire  Gage,"  but  it  should  not  be  confused  with  the  Stubs' 
Steel  Wire  Gage. 

British  Standard  Wire  Gage.  This  gage,  usually  called  simply  the  * 'Standard  Wire 
Gage"  and  abbreviated  "S.W.G.,"  is  also  known  as  the  "New  British  Standard"  (ab- 
breviated "N.B.S."),  the  English  Legal  Standard,  or  the  Imperial  Wire  Gage,  and  is  the 
legal  standard  of  Great  Britain  for  all  wires,  as  fixed  by  order  in  Council,  August  23,  1883. 
It  was  constructed  by  modifying  the  Birmingham  Wire  Gage,  so  that  the  differences  be- 
tween successive  diameters  were  the  same  for  short  ranges,  i.e.,  so  that  a  graph  representing 
the  diameters  consists  of  a  series  of  a  few  straight  lines. 

Edison  Wire  Gage.  The  size  of  a  wire  on  this  gage  is  equal  to  its  cross-sectional  area 
in  circular  mils  divided  by  1000.  For  example,  a  solid  wire  0.2  in.  in  diameter  has  the 
number  (200)2/1000  =  40.  This  gage  is  now  rarely  used. 

Metric  Wire  Gage.     The  gage  number  is  ten  times  the  diameter  in  millimeters. 

Other  Gages.  In  addition  wire  sizes  are  sometimes  specified  in  terms  of  the  "Old 
English  Wire  Gage,"  known  also  as  the  "London  Gage,"  and  the  "Stubs'  Steel  Wire 
Gage."  The  Old  English  Wire  Gage  is  the  same  as  B.W.G.  for  all  gage  numbers  under  20. 

Comparison  of  Wire  Gages.  A  comparison  of  the  different  gages,  in  terms  of  the  diam- 
eters (in  mils  or  thousandths  of  an  inch)  of  solid  wires  corresponding  to  the  various  num- 
bers, is  given  in  Table  38.  The  cross-section  in  circular  mils  is  the  square  of  the  diameter 
in  mils. 


1-70  MATHEMATICS,  UNITS,  AND  SYMBOLS 

Table  38.    Comparison  of  Wire  Gage  Diameters  in  Mils 

(Bureau  of  Standards,  Circulars  31  and  67) 


Gage 
No. 

America 
wire 
gage 
(B.  &S 

Steel 
wire 
gage 

Birming 
ham  wir 
gage 
(Stubs' 

Old 

English 
wire  gag 
(London 

Stubs' 
steel 
wire 
gage 

(British 
Standarc 
wire 
gage 

Metric 
gage  * 

Gage 

No. 

7-0 

6-0 
5-0 
4-0 
3-0 
2-0 
0 
1 
2 
3 
4 
5 
6 
7 
8 
9 
10 
11 
12 
13 
14 
15 
16 
17 
18 
19 
20 
21 
22 
23 
24 
25 
26 
27 
28 
29 
30 
31 
32 
33 
34 
35 
36 
37 
38 
39 
40 
41 
42 
43 
44 
45 
46 
47 
48 
49 
50 

490.0 
461.5 
430.5 
393.8 
362.5 
331.0 
306.5 
283.0 
262.5 
243.7 
225.3 
207.0 
192.0 
177.0 
162.0 
148.3 
135.0 
120.5 
105.5 
91.5 
80.0 
72.0 
62.5 
54.0 
47.5 
41.0 
34.8 
31.7 
28.6 
25.8 
23.0 
20.4 
18.1 
17.3 
16.2 
15.0 
14.0 
13.2 
12.8 
11.8 
10.4 
9.5 
9.0 
8.5 
8.0 
7.5 
7.0 
6.6 

500 
464 
432 
400 
372 
348 
324 
300 
276 
252 
232 
212 
192 
176 
160 
144 
128 
116 
104 
92 
80 
72 
64 
56 
48 
40 
36 
32 
28 
24 
22 
20 
18 
16.4 
14.8 
13.6 
12.4 
11.6 
10.8 
10.0 
9.2 
8.4 
7.6 
6.8 
6.0 
5.2 
4.8 
4.4 
4.0 
3.6 
3.2 
2.8 
2.4 
2.0 
1.6 
1.2 

7-0 
6-0 
5-0 
4-0 
3-0 
2-0 
0 
1 
2 
3 
4 
5 
6 
7 
8 
9 
10 
11 

12 
13 
14 
15 
16 
17 
18 
19 
20 
21 
22 
23 
24 
25 
26 
27 
28 
29 
30 
31 
32 
33 
34 
35 
36 
37 
38 

39 
40 
41 
42 
43 
44 
45 
46 
47 
48 
49 

460 
410 
365 
325 
289 
258 
229 
204 
182 
162 
144 
128 
114 
102 
91 
81 
72 
64 
57 
51 
45 
40 
36 
32 
28.5 
25.3 
22.6 
20.1 
17.9 
15.9 
14.2 
12.6 
11.3 
10.0 
8.9 
8.0 
7.1 
6.3 
5.6 
5.0 
4.5 
4.0 
3.5 
3.1 

454 
425 
380 
340 
300 
284 
259 
238 
220 
203 
180 
165 
148 
134 
120 
109 
95 
83 
72 
65 
58 
49 
42 
35 
32 
28 
25 
22 
20 
18 
16 
14 
13 
12 
10 
9 
8 
7 
5 
4 

454 
425 
380 
340 
300 
284 
259 
238 
220 
203 
180 
165 
148 
134 
120 
109 
95 
83 
72 
65 
58 
49 
40 
35 
31.5 
29.5 
27.0 
25.0 
23.0 
20.5 
18.75 
16.50 
15.50 
13.75 
12.25 
11.25 
10.25 
9.50 
9.00 

7.50 
6.50 
5.75 
5.00 
4.50 

227 
219 
212 
207 
204 
201 
199 
197 
194 
191 
188 
185 
182 
180 
178 
175 
172 
168 
164 
161 
157 
155 
153 
151 
148 
146 
143 
139 
134 
127 
120 
115 
112 
110 
108 
106 
103 
101 

99 
97 
95 
92 
88 
85 
81 
79 
77 
75 
72 

3.94 
7.87 
11.8 
15.7 
19.7 
23.6 
27.6 
31.5 
35.4 
39.4 

47.2 

55.1 

63.0 

70.9 

78.7 

98.4 

118 

138 

157 

6.2 

6.0 

5.5 

177 

5.2 

4.8 
4.6 

A    A 

*  For  diameters  corresponding  to  metric  ca 
those  of  12,  14,  etc.,  by  tea. 

ov 
ge  numbers,  1.2,  1.4,  ] 

1.0 
.6,  1.8,  2.5 

197 
3.5,  and  4 

.5,  divide 

ABBREVIATIONS  FOR  ENGINEERING  TERMS 


1-71 


SYMBOLS  AND  ABBREVIATIONS 


19.  ABBREVIATIONS  FOR  ENGINEERING  TERMS 


NOTE:  This  list  is  a  selection  of  American  Tentative  Standard  abbreviations,  for  scientific  and 
engineering  terms,  recommended  by  the  American  Standards  Association.     (See  ASA,  Z10.1 — 1941.) 


Absolute abs 

Acre spell  out 

Alternating-current  (as  adjective) ac 

Ampere amp 

Ampere-hour amp-hr 

Angstrom  unit A 

Atmosphere atm 

Atomic  weight at.  wt. 

Average avg 

Avoirdupois avdp 

Barometer bar. 

Barrel bbl 

Baume1 Be" 

Boiler  pressure spell  out 

Boiling  point bp 

Brake  horsepower bhp 

Brake  horsepower-hour bhp-hr 

Brinell  hardness  number Bhn 

British  thermal  unit Btu  or  B 

Calorie cal 

Candle c 

Candle-hour c-hr 

Candlepower cp 

Centigram eg 

Centiliter cl 

Centimeter cm 

Centimeter-gram-second  (system) cgs 

Chemically  pure '. cp 

Circular cir 

Circular  mils cir  mils 

Coefficient coef 

Cologarithm colog 

Concentrate cone 

Conductivity cond 

Constant , const 

Cord cd 

Cosecant esc 

Cosine cos 

Cotangent cot 

Coulomb spell  out 

Counter  electromotive  force cemf 

Cubic cu 

Cubic  centimeter cu  crn,  cm3,  cc 

Cubic  feet  per  minute cfm 

Cubic  foot cu  f t 

Cubic  inch cu  in. 

Cubic  meter cu  m  or  m3 

Cubic  yard cu  yd 

Cycles  per  second spell  out  or  c 

Decibel db 

Degree deg  or  ° 

Degree  Centigrade C 

Degree  Fahrenheit F 

Degree  Kelvin K 

Degree  Reaumur R 

Diameter diam 

Direct-current  (as  adjective) d-c 

Dozen doz 

Dram dr 

Efficiency eff 

Electric elec 

Electromotive  force emf 


Equation eq 

External ext 

Farad spell  out  or  f 

Foot ft 

Foot-candle f t-c 

Foot-Lambert f t-L 

Foot-pound f t-lb 

Foot-pound-second  (system) fps 

Freezing  point fp 

Fusion  point fnp 

Gallon gal 

Grain spell  out 

Gram g 

Gram-calorie g-cal 

Henry h 

Horsepower hp 

Horsepower-hour hp-hr 

Hour hr 

Hundred C 

Hyperbolic  sine sinh 

Hyperbolic  cosine cosh 

Hyperbolic  tangent tanh 

Inch in. 

Inch-pound in-lb 

Internal int 

Joule j 

Kilocycles  per  second kc 

Kilogram kg 

Kilogram-calorie kg-cal 

Kilogram-meter kg-m 

Kiloliter Id 

Kilometer km 

Kilovolt kv 

Kilovolt-ampere kva 

Kilowatt kw 

Kilowatthour kwhr 

Lambert L 

Latitude lat  or  <f> 

Linear  foot lin  ft 

Liter 1 

Liquid liq 

Logarithm  (common) log 

Logarithm  (natural) loge  or  hi 

Longitude long,  or  X 

Lumen 1 

Lumen-hour 1-hr 

Magnetomotive  force mmf 

Maximum max 

Melting  point mp 

Meter m 

Meter-kilogram m-kg 

Mho spell  out 

Microampere Ma  or  mu  a 

Microfarad /if 

Micromicron ij.fi  or  mu  mu 

Micron M  or  mu 

Microwatt MW  or  mu  w 

Mile spell  out 

Milliampere ma 

Milligram mg 

Millihenry , mh 


1-72 


MATHEMATICS,   UNITS,  AND  SYMBOLS 


Abbreviations  for  Engineering  Terms— Continued 

Milliliter ml 

Millimeter mm 

Millimicron 

Million 

Millivolt P™ 

Mean  horizontal  eandlepower mncp 

Miles  per  hour , mPh 

Minimum • 

Minute • 

Minute  (angular  measure) 

Qkm  spell  out  or  £2 

Ounce 

Ounce-foot 

Ounce-inch 


or  m  mu 
sPeU  out 


mm 
min 


oz'ltl 
oz-in- 


Pint 


Potential  ...........................   speU  out 

Pound  ..................................  •  £ 

Pound-foot  ............................  Tilb:±t; 

Pound-inch  ...........................  lb-m- 

Pounds  per  square  foot  ...................  psi 

Pounds  per  square  inch  ...................   PS1 

Power  factor  ..................   spell  out  or  pf 

Quart  ...................................   <# 

Radian  ............................  sPeU  out 

Reactive  kilovolt-ampere  ................  kvar 

Reactive  volt-ampere  ....................   var 

Revolutions  per  minute  ..................  rPm 

Revolutions  per  second  ...................   rPs 

Root  mean  square  .......................  rms 


see 
sec 


SP 


Secant 

Second  ............ 

Second  (angular  measure) 

Sine 

Specific  gravity 

Specific  heat  ........................... 

Spherical  candlepower  ....................  SCP 

Square  ..................................   acj 

Square  centimeter  ..............   sq  cm  or  cm2 

Square  foot  ............................   8flft 

Square  inch  .......................  ••••    S(J  m£ 

Square  kilometer    ..............  «Q  ^m  or  km 

Square  meter  .....................    Bq  in  or  m< 

Square  micron  ............   sq  ^  or  sq  mu  or  ^ 

Square  root  of  mean  square  ...............   rms 

Standard  ..............................   sfcd 

Tangent  ...............................  •    tan 

Temperature  ..........................    temP 

Thousand  ..............................  •   M 

Ton  ...............................   8Pe11  oufc 

Versed  sine  .............................   VGrs 

Volt  .....................................    v 

Volt-ampere  .............................   va 

Watt  ....................................   w 

Watthour  ..............................  whr 

Weight  .................................   wt 

Yard  ...................................   3rd 

Year  ....................................   V* 


20.  LETTER  SYMBOLS  FOR  THE  MAGNITUDES  OF 
ELECTRICAL  QUANTITIES 

(Tentative  American  Standard  Z10.5-1947)  t 
In  the  alphabetical  order  of  the  names  of  the  quantities 

Each  quantity  appears  at  only  one  place  in  this  table  (with  a  few  exceptions),  listed 
alphabetically  under  its  preferred  name.  The  non-preferred  names  appear  in  parentheses 
under  the  preferred  names.  Deprecated  names  are  also  in  parentheses  and  in  addition 
are  asterisked  thus:  (electric  force)*. 

Names  beginning  with  the  qualifying  adjectives,  electric,  electrostatic,  dielectric,  magnetic, 
mutual,  self,  and  relative,  are  listed  under  the  term  that  is  so  qualified. 

Symbols  for  scalar  quantities,  whose  values  are  expressed  by  real  numbers,  are  printed 
in  ordinary-face  italic  letters. 

Symbols  for  vector  quantities  are  printed  in  "bold-face  Roman  letters. 

Symbols  for  phasor  quantities,  whose  values  are  expressed  by  complex  numbers,  are 
printed  in  bold-face  italic  letters. 


Item 

Quantity 

Symbol 

Item 

Quantity 

Symbol 

1 

admittance 

y 

7 

line  d.  of  charge 

X 

2 

attenuation  constant 

a 

8 

surface  d,  of  charge 

<r 

3 

capacitance 

C 

9 

volume  d.  of  charge 

P 

(capacity)  * 

10 

conductance 

O 

(permittance)  * 

11 

conductivity 

y 

4 

capacitivity 

6 

12 

conductivity, 

A 

dielectric  constant 

equivalent 

(permittivity)  * 

13 

coupling  coefficient 

k 

of  evacuated  space 

«T 

14 

current 

I 

5 

capacitivity,  relative 

*r 

(intensity  of  current)  * 

relative  dielectric  constant 

15 

current  density 

(specific  inductive  capacity)     * 

16 

sheet  c.d.  (linear  c.d.) 

A 

6 

charge,  electric  or  quantity  of 

Q 

17 

damping  constant  or  coefficient 

8 

electricity 

(decay  constant) 

charge  density 

*  Deprecated  name. 


f  Reprinted  by  permission  of  the  American  Institute  of  Electrical  Engineers. 


LETTER  SYMBOLS  FOR  ELECTRICAL  QUANTITIES      1-73 


Letter  Symbols  for  the  Magnitudes  of  Electrical  Quantities — Continued 


Item 

Quantity 

Symbol 

Item 

Quantity 

Symbol 

18 

dielectric  constant 

£ 

50 

phase  constant 

(3 

see  capacitivity 

wavelength  constant 

dielectric,  a  qualifier 

(wave  number) 

see  term  that  it  qualifies 

51 

polarization,  electric 

19 

displacement,  electric 

D 

52 

polarization,  magnetic 

Bi 

20 

efficiency 

f] 

intrinsic  induction 

21 

elastance 

S 

metallic  induction 

mutual  e.    Sm,  Src 

53 

pole  strength 

m 

self  e.     8,  Soc 

54 

potential,  electric 

V 

22 

elastivity 

ff 

(electromagnetic  scalar  p.) 

electric,  a  qualifier 

55 

potential,  retarded  scalar 

see  term  that  it  qualifies 

56 

potential,  magnetic 

M,& 

23 

electronic  charge 

e 

(magnetic  scalar  p.) 

(absolute  value  of) 

m.  pot.  difference 

24 

electromotive  force 

E 

57 

potential,  magnetic  vector  p. 

A 

(electromotance) 

58 

potential,  retarded  vector  p. 

A.r 

(potential  difference,  electric) 

59 

power,  active 

P 

(voltage)  * 

60 

power,  reactive 

Q 

25 

energy 

W 

volt-amperes,  reactive 

26 

force 

F 

61 

power,  apparent 

S 

27 

flux,  displacement  f. 

^ 

volt-amperes 

(flux  of  e.  displacement) 

62 

power  factor 

Fp 

28 

flux,  magnetic 

$ 

63 

propagation  constant 

7 

(flux  of  magnetic  induction) 

64 

Poynting  vector 

1  n 

29 

flux-linkage 

65 

quantity  of  electricity 

Q 

30 

frequency 

/ 

charge,  electric 

'31 

frequency,  angular 

CO 

66 

quality  factor  of  a  reactor 

Q 

angular  velocity 

figure  of  merit  of  a  reactor 

32 

frequency,  rotational 

n 

67 

reactance 

X    , 

33 

impedance 

Z 

capacitative  r. 

Xc 

mutual  i.    Zm,  Zrc 

inductive  r. 

XL 

self  i.     Z,  Zcc 

mutual  r.    %m,  Xrc 

34 

induction,  magnetic 

B 

selfr.     Z,  Zcc 

(magnetic  flux  density) 

68 

reactive  factor 

Fq 

35 

inductance 

L 

69 

reluctance 

(R 

mutual  i.    Lm,  Lrc 

70 

reluctivity 

v 

self  i.    L,  LCC 

71 

resistance 

R 

36 

intensity,  electric 

E,K 

mutual  r.     Rm,  Rrc 

(electric  field  intensity) 

self  r.    R,  Rcc 

(electric  field  strength) 

72 

resistivity 

P 

(electric  force)  * 

73 

resistance-temperature  coefficient 

a 

(electric  field)  * 

rotative  operators 

37 

intensity,  magnetic  or  magnetiz- 

H 

74 

90°,  V-  * 

j 

ing  force 

75 

120°,  -\/~\ 

a 

(magnetic  field  strength) 

self,  a  qualifier 

(magnetic  force)  * 
magnetic,  a  qualifier 

76 

see  term  that  it  qualifies 
slip 

see  term  that  it  qualifies 

77 

susceptance 

S 

38 

magnetomotive  force 

M.JF 

susceptibility 

(m.  potential  difference) 

78 

dielectric  s. 

y 

39 

magnetomotance 
moment,  electric 

P 

79 

intrinsic  capacitivity 
magnetic  s. 

K 

40 
41 
42 

moment,  magnetic 
number  of  conductors  or  turns 
number  of  poles 

m 

N 
P 

80 
81 

intrinsic  permeability 
symmetrical  components  (Note  5) 
temperature 

43 

number  of  phases 

m 

T,  (0) 

44 

period 

T 

82 

time 

i 

45 

permeance 

<P,  A 

83 

time  constant 

T 

46 

permeability,  magnetic 
of  evacuated  space 

to 

84 

85 

velocity  of  light 
vibration  constant 

c 
P 

47 
48 

permeability,  relative 
(permittivity)  *  (see  capacitivity) 

Mr 

86 

(oscillation  constant) 
wavelength 

X 

49 

phase  angle 

9 

87 

wavelength  constant 

phase  constant 

88 

work 

W 

*  Deprecated  name. 


1-74  MATHEMATICS,   UNITS,   AND   SYMBOLS 

Note  1.     Designation  of  maximum,  instantaneous,  rms,  and  average  values. 

Where  distinctions  between  maximum,  instantaneous,  root-mean-square  (effective),  and 
average  values  are  necessary,  Emt  Im,  Qw,  and  Pm  are  recommended  for  maximum  values; 
e,  t,  #,  and  p  for  instantaneous  values,  E,  I,  and  Q  for  root-mean-square  values  and  £,a,  /0, 
Qa,  and  P  for  average  values. 
Note  2.     Quantities  per  unit  volume,  area,  or  length. 

It  is  recommended  that  quantities  per  unit  volume,  area,  length,  etc.,  be  represented  as 
far  as  practicable  by  lower-case  letters  corresponding  to  the  cap  letters  which  represent  the 
total  quantities,  or  by  the  cap  letters  with  the  subscript  1,  except  for  those  quantities  for 
which  this  table  has  symbols  for  the  quantity  per  unit  volume,  area,  etc. 
Note  3.     Distinction  between  the  symbols  V  and  E  for  potential  and  electromotive  force. 

The  distinction  between  the  use  of  V  for  potential  and  E  for  electromotive  force  is  : 

V  is  to  be  used  for  potentials  or  potential  differences  that  are  attributed  solely  to  that 
distribution  of  electric  field  intensities  which  is  computed  (by  the  inverse  square  law  of 
force)  from  the  segregated  charges  of  the  field. 

E  is  to  be  used  for  the  emf  along  a  path  from  a  terminal  A  to  a  terminal  B  when  in  the 
region  A  to  B  one  or  more  non-electrostatic  types  of  electric  intensities  exist,  or  turbulent 
actions  occur  —  as  in  voltaic  cells,  electrostatic  generators,  and  electromagnetic  sources  of 
emf. 

Note  4.     The  sequence  of  the  double  subscripts  to  multiplying  operators. 

The  sequence  of  the  double  subscripts  to  the  multiplying  operators  (mutual  impedances, 
resistances,  or  elastances  or  transconductances,  etc.)  that  occur  in  the  fundamental  equa- 
tions of  networks  is  to  be  determined  by  the  following  consideration  : 

Consideration.  The  set  of  fundamental  equations  (e.g.,  Kirchhoff's  emf  equations) 
should  yield  a  determinant  in  which  the  subscript  sequence  conforms  to  the  mathe- 
matician's convention  for  writing  determinants;  namely, 

Convention.  In  the  double  subscripts  of  the  elements  of  a  determinant,  the  subnumbor 
designating  the  "row"  is  to  precede  the  subnumber  designating  the  "column"  to  which  the 
element  belongs,  or  the  order  is  erc-  Thus 


This  consideration  leads  to  the  following  rule  : 
Rule  for  writing  double  subscripts. 

The  first  subnumber  in  the  symbol  for  a  multiplying  operator  designates  the  number  of 
the  circuit  in  which  the  product  of  the  multiplication  is  measurable,  while  the  second  sub- 
number  designates  the  number  of  the  circuit  in  which  the  operand  or  multiplicand  is 
measurable. 

As  an  illustration,  Eirchhoff's  emf  law  for  the  emfs  of  the  rth  circuit  due  to  the  currents 
in  all  the  circuits  of  a  network  is  written: 


Er,d  = 

(ET,d  being  the  driving  emf  impressed  in  the  rth  circuit)  . 
Note  5.     Notation  for  symmetrical  components. 

The  standard  notation  for  designating  the  symmetrical  components  of  the  currents  and 
potential  differences  in  unbalanced  polyphase  systems  is  that  subscript  notation  in  which: 

(a)  double  subscripts  are  added  to  the  symbols  for  current  and  potential  difference; 

(b)  the  first  and  second  subscripts  designate,  respectively,  the  phase  and  the  sequence  to 
which  the  component  belongs; 

(c)  the  first,  or  phase,  subscript  may  be  the  phase  number,  or  the  phase  letter,  or  a  two- 
letter  combination  that  designates  (on  a  diagram)  both  the  phase  and  the  direction  in 
the  phase; 

(d)  the  second  or  sequence,  subscript  is  always  to  be  the  number  that  designates  the 
sequence  to  which  the  component  belongs;  the  positive,  negative,  and  zero  sequence  com- 
ponents in  three-phase  systems  being  designated  by  the  numbers  1,  2,  and  0,  respectively. 
Illustration  of  notation: 

la,   ~   I  al  +  -Fo2  +  IaQ 
2b  —  Ibl  +  2  b2  +  IbQ 

Ic  =  la.  + 


1-76 


MATHEMATICS,  UNITS,  AND   SYMBOLS 


21.  STANDARD  GRAPHICAL  SYMBOLS 

(Approved  by  American  Standards  Association,  Nov.  1,  1942) 
(Revised  by  American  War  Standard,  April  18,  1944) 


1.  Ammeter 
2.  Antenna 

3.  Antenna,  Loop 

-/A\- 

LM 

0 

17.  Jack 
18.  Key 
19.  Lightning  Arrester 

20.  Loudspeaker 

o  *    "Vf] 

i 

=oC] 

4.  Arc 

1<: 

21.  Microphone 

=3 

5.  Battery 

Long  line  always  positive 
but  polarity  may  be  Indicated 
In  addition. 

H^ 

22.  Phototube 

$ 

6.  Capacitor,  Fixed                            —  L- 
Condenser,  Fixed                          """P* 

The  curved  electrode  Identifies 
the  outermost  electrode  where  applicable 
or  the  negative  electrode  for  electrolytic 
capacitors. 

7.  Ca-pacitor,  Fixed,  Shielded        —  -}  (  

23.  Piezoelectric  Plate 
24.  Resistor 

25.  Resistor,  Adjustable  or 
Variable 

-0- 

-AAA/V 

8.  Capacitor,  Variable 

The  curved  portion  is  the 
movable  electrode. 

9.  Capacitor,  Variable,  Shielded 

4L 

26.  Spark  Gap,  Plain 
27.  Spark  Gap,  Quenched 
28.  Spark  Gap,  Rotary 

-D  0- 

-niniiih 

_  AA  

10.  Counterpoise 

rh 

29.  Telephone  Receiver 
30.  Telephone  Transmitter 

cjcp 
=OJ 

U.  Crystal  Detector 
12.  Galvanometer 

•f 

31.  Thermoelement 
32.  Transformer,  Air  Core 

_LL 

13.  Ground                                         J=- 
14.  Inductor                                 -^rjfftftfX- 

15.  Inductor,  Adjustable  or  Variable 

—  onooo  —  . 

33.  Transformer,  Iron  Core. 

3C 

34.  Transformer,  with  Variable  Coupling 

iff. 

16.  Inductor  Iron  Core 

£__  l 
-^Tnj^cr^ 

35.  Voltmeter 
36.  Wires,  Crossed,  not  joined 

»     *  >  i  my 

+ 

-jam- 

37.  Wires,  Joined 

_ru_ 

STANDARD  GRAPHICAL  SYMBOLS 


l-'J 


1.  Anode  or  Plate 

(Including  Collector) 


2.  Cathode-Ray  Tube  wfifh 
Electrostatic  Deflection 


3.  Cathode-Ray  Tube  for 
Magnetic  Deflection 


4.  Cold  Cathode 

(Including  Ionic-Heated  Caihode) 

5.  Deflecting,  Reflecting,  or 
Repelling  Electrode 

(Electrostatic  Type) 


6.  Diode 

(Cold-cathode  and 
Gas  Content) 


7.  Directly  Heated  Cathode 

(Filament  Type) 


8.  Double-Cavity  Resonator 
Eav.eJope 


9.  Double-Cavity  Velocity- 
Modulation  Tube  with 
Collecting  Electrode 


»10.  Dynod'e 


11.  Excitor 

(Contacto'f'T.ype) 


Electron  Tubes 

(ASA  232.10-1944) 


14.  Beater 


12.  Gas-Filled  Envelope 


Located  as      X^^X 
convenient     f- >+  \ 


13.  Grid 

(including  Beam-Confining  or 
Beam-Forming  ELectnodes) 


n 


15.  High-Vacuum  Envelope 


16.  Ignitor 


Ignltor 


17.  Indirectly  Heated  Cathode 


18.  Ionic-Heated  Cathode  with 
Supplementary  Heater 

19.  Loop  Coupling 

(Electromagnetic  Type) 


20.  Mercury  Pool  Tube  with 
Excitor,  Control  Grid,  and 
Holding  Anode 


21.  Mercury  Pool  Tube  with 
Ignitor  and  Control  Grid 


22.  Pentode 

(Suppressor  or  Beam- 
confining  Electrodes) 


23.  Photoelectric  Cathojde 


24.  Phototube 


25.  Phototube 

(Multiplier  Type) 


Y 


1-78  MATHEMATICS,  UNITS,  AND  SYMBOLS 

Electron  Tubes — Continued 
(ASA  Z32.10-1944) 


26.  Pool  Cathode 

27.  Resonant  Magnetron 

28.  Shield  within  Envelope  ( i 

External  Connection 


29.  Single-Cavity  Resonator 
Envelope 


30.  Single-Cavity  Velocity- 
Modulation  Tube  with 
Reflecting  Electrode 


31.  Target,  X-Ray 


32.  Transit-Time  Split-Plats  Type 
Magnetron  with  Stabilizing 
Deflecting  Electrodes  and 
Internal  Circuit 


33.  Triode  with  Filamentary 
Cathode 


34.  Triode  with  Indirectly 
Heated  Cathode  and 
Envelope  Connection 


35.  Triode  with  Indirectly 
Heated  Cathode  and 
Envelope  Connected 
to  Base  Terminal 


36,  Triode-Heptode  with  Rigid 
Envelope  Connection 


Small  Pin 

37.  Tube  Base  Terminals 

Large  Pin 

Rigid  Terminals 
Flexible  Leads. 

38.  Tube  Envelope  Terminals 


39.  This  figure  illustrates 
how   tube    symbols 
may  be  placed  in  any 
convenient  position 
as  shown  in   a 
communication 
transformer  circuit. 


40.  General  Notes 

(a)  The  diagram  for  a  tube  having  more  than 
one  heater  shall  show  only  one  heater  symbol 
(inverted  V)  unless  the  heaters  have  entirely 
separate  connections.  If  a  tap  Is  made,  one  heater 
symbol  shall  still  be  shown,  and  the  tap  shall  be 
shown  at  the  vertex  of  the  heater  symbol,  regardless 
of  the  actual  division  of  voltage  across  the  heater 

(6)  Item  (a)  shall  apply  also  to  filaments.  In 
case  of  a  tap,  either  brought  out  to  a  pin  connection 
or  internally  connected  as  to  a  suppressor  grid,  the 
tap  shall  be  shown  at  the  vertex  of  the  filament 
symbol,  regardless  of  the  actual  division  of  voltage 
across  the  filament. 

(c)  A  type  having  more  than  one  cathode  shall 
be  shown  as  having  a  single  cathode  unless  separate 
cathode  connections  are  made. 

(d)  A  type  having  two  or  more  grids  tied 
Internally  shall  be  shown  with  symbols  for  each  grid, 
except  when  the  grids  are  adjacent  in  the  tube 
structure.  Thus  the  diagram  for  a  twin  pentode 
having  a  common  screen-grid  connection  for  each 
section  and  for  a  converter  tube  having  the  No.  3 
and  the  No.  5  grids  connected  Internally  will  show 
separate  symbols  for  each  grid.  However,  a  trlode 
where  the  control  grid  Is  physically  In  the  form  of 
two  grid  windings  would  show  only  one  grid. 

(e)  A  type  having  a  grid  adjacent  to  a  plate  but 
Internally  connected  to  the  plate  to  form  a  portion  of 
it  shall  be  shown  as  having  a  plate  only, 

(/)  Associated  parts  of  a  circuit  such  as  deflecting 
coils,  field  coils,  etc.,  are  not  a  part  of  the  tube 
symbol  but  may  be  added  to  the  circuit  in  the  form 
of  standard  symbols  as  shown  in  ASA  Z32.3  or  ASA 
Z32.5.  For  example,  resonant-type  magnetron  plus 
symbol  for  ferromagnetic  inductor  would  be  shown 


r) 

V 


Uj&ly — 


PRINCIPAL  PHYSICAL  CONSTANTS   AND   RATIOS       1-79 


22.  USE  OF  GREEK  ALPHABET  FOR  SYMBOLS 


Capital 

Lower 
Case 

Name 

Commonly  Used  to  Designate 

A 

a 

Alpha 

Angles.    Area.    Coefficients.    Attenuation  constant. 

B 

JS 

Beta 

Angles.    Flux  density.    Coefficients. 

r 

7 

Gamma 

Conductivity.    Specific  gravity.    Propagation  constant. 

A 

8 

Delta 

Variation.    Density.    Damping  coefficient. 

E 

€ 

Bpsilon 

Base  of  natural  logarithms.    Capacitivity. 

z 

r 

Zeta 

Impedance.    Coefficients.    Coordinates. 

H 

t\ 

Eta 

Hysteresis  coefficient.    Efficiency. 

e 

e 

Theta 

Temperature.    Phase  angle. 

I 

i 

Iota 

K 

» 

Kappa 

Dielectric  constant.    Susceptibility. 

A 

X 

Lambda 

Wavelength. 

M 

M 

Mu 

Micro.    Amplification  factor.    Permeability. 

N 

V 

Nu 

Reluctivity. 

3 

£ 

Xi 

0 

0 

Omicron 

n 

TT 

Pi 

Ratio  of  circumference  to  diameter  =  3.  1  4  1  6. 

p 

P 

Rho 

Resistivity. 

s 

ff 

Sigma 

Capital:  sign  of  summation. 

T 

T 

Tau 

Time  constant.    Time  phase  displacement. 

T 

U 

Upsilon 

$ 

$  or  <p 

Phi 

Magnetic  flux.    Angles. 

X 

X 

Chi 

* 

t 

Psi 

Dielectric  flux.    Phase  difference. 

ft 

CO 

Omega 

Capital:  ohms.    Lower  case:  angular  velocity,  or  2tr  X  frequency. 

CONSTANTS 

By  Carl  C.  Chambers 

23.  PRINCIPAL  PHYSICAL  CONSTANTS  AND  RATIOS  * 


Velocity  of  light 

Ratio  of  electrostatic  to  electromagnetic  units . 

Volume  of  a  perfect  gas  (0  deg  cent  and  normal 

atmospheric  pressure) 

Normal  atmospheric  pressure 

45  deg  cent  atmospheric  pressure 

Ice  point  (absolute  scale) 

Mechanical  equivalent  of  heat  (15  deg  cent) . . 
Electrical  equivalent  of  heat  (15  deg  cent) .... 

Faraday  constant 

Electronic  charge 

Planck  constant 

Acceleration  of  gravity 

Electrochemical  equivalent  of  silver 

Wave  length  of  red  cadmium  line  (15  deg  cent, 
normal  atmospheric  pressure) 

Effective  grating  space  of  calcite  (18  deg  cent) 

Avogadro's  number 

Boltzmann  constant 

Stefan-Boltzmann  constant 

Mass  of  the  electron 

Ratio  of  mass  of  H  to  mass  of  electron  (meas- 
ured by  deflection) 


(2.99776  db  0.00004)  X  1010  cm  sec"1 


f     (2.9971  ±  0.0001)  X  1010  < 


c"^ 


(in 


\  (2.9978  ±  0.0001)  X  1010  cm  sec"1  (in  absolute  units) 

(22.4146  ±  0.0006)  X  103  cm3  mole"1 
(1.013246  db  0.000004)  X  106  dynes  cm"2 
(1.013195  ±  0.000004)  X  106  dynes  cm"2 

273.18  ±  0.01°  K 

4.1855  ±  0.0004  abs  joule  cal"1 

4.1847  ±  0.0003  int  joule  cal"^ 

96494  ±  5  int  coulombs  g-equiv"1 

(4.8025  ±  0.0010)  X  10~10  abs-es  unit 

(1.60203  ±  0.00034)  X  10"20  abs-em  unit 

(6.624  ±  0.002)  X  10"27  erg  sec 

980.665  cm  sec"2 
1.11800  X  10~3  g.  int  coulombs"1 

6438.4696  I.A.  f 

3.02904  X  10~8  cm 

(6.0228  ±  0.0011)  X  1023  mole"1 

(1.3708  ±  0.0014)  X  10"16  erg  deg"1 

(5.672  ±  0.003)  X  10~5  erg  cm-2  deg~4  sec"1 

(9.1066  ±0.0032)  X  KT28  g 

1837.5  ±  0.5 


*  Values  taken  from  Birge,  Rev.  of  Mod.  Phys.,  Vol.  13,  No.  4  (October,  1941). 
t  This  defines  the  international  angstrom  unit  (I. A.).    The  unit  is  of  the  order  of  1  part  in  several 
million  different  from  10~8  cm. 


1-80 


MATHEMATICS;    UNITS,    AND    SYMBOLS 


24.  STANDARD  RADIO-FREQUENCY  RANGES 

The  International  Telecommunication  Union  in  1947  adopted  officially  the  Nomen- 
clature of  Frequencies  shown  in  Table  39.  The  designation  of  each  metric  subdivision  of 
wavelength  range  is  the  name  of  the  metric  unit  of  length  which  is  equal  to  the  shortest 
wavelength  in  the  range.  The  range  number  is  the  power  of  10  which  represents  the 
approximate  mean  frequency  of  that  range.  The  frequency  subdivision  designations  are 
those  first  adopted  by  the  Armed  Forces  of  the  United  States  and  subsequently  by  other 
branches  of  the  government. 

The  table  is  based  on  the  approximation  that  the  wave  velocity  is  300,000,000  meters 
per  second.  In  any  case,  where  the  required  precision  makes  this  assumption  inadequate, 
the  exact  boundaries  of  the  ranges  should  be  based  on  the  frequency  range,  not  the  wave- 
length range.  .,  .,•  ,  *  • 

It  is  suggested  that  the  term  radio  be  added  where  there  is  any  possibility  of  confusion. 
A  power  engineer  might  be  considerably  startled  to  hear  100  kc  referred  to  as  a  'low 
frequency." 

The  term  microwave  has  been  variously  used:  (1)  as  referring  to  waves  less  than  1  meter 
in  length;  (2)  as  certainly  including  ranges  10  and  11,  and  part  of  9;  (3)  as  referring  to 
waves  using  cavities  instead  of  LC  tuned  circuits;  and  (4)  as  referring  to  waves  transmitted 
by  wave  guides.  Where  any  confusion  is  possible,  the  term  should  be  eschewed  or  clarify- 
ing text  added. 

Table  39.     Nomenclature  of  Frequencies 


Range 

Frequency  Range 

Wavelength 
Range 

"Mum 

Frequency 

J.NUm- 

her 

Lower 

Upper 

Lower 

Upper 

Subdivision 

Subdivision 

N 

Limit 

Limit 

Limit 

Limit 

(Inch) 

(Excl.) 

(Excl.) 

CIncl.) 

0 

0.3  c 

3c 

I 

3c 

30  c 

2 

30  c 

300  c 

3 

300  c 

3kc 

4 

3kc 

30  kc 

10km 

100km 

Myriametric  waves 

VLF  (very  low  frequency) 

5 

30  kc 

300  kc 

1km 

10km 

Kilometric  waves 

LF  (low  frequency) 

6 

300  kc 

3,000fcc 

Ihm 

lOhm 

Hectometric  waves 

MF  (medium  frequency) 

7 

3,000kc 

30,000  kc 

1  dkm 

10  dkm 

Decametric  waves 

HF  (high  frequency) 

8 

30,000kc 

300  Me 

1  m 

10m 

Metric  waves 

VHF     (very     high     fre- 

quency) 

9 

300  Me 

3,  000  Me 

1  dm 

10dm 

Decimetric  waves 

UHF     (ultra     high     fre- 

quency) 

10 

3,000  Me 

30,  000  Me 

1  cm 

10cm 

Centimetric  waves 

SHF     (super     high     fre- 

quency) 

11 

30,000  Me 

300  kMc 

1  mm 

10mm 

Millimetric  waves 

EHF  (extremely  high  fre- 

quency) 

12 

300  kMc 

3,000  kMc 

13 

3,000  kMc 

30,000  kMc 

14 

30,  000  kMc 

300  MMc 

Note  I .     Ranges  0-3  and  12-14  not  standardized  by  I.T.U. 

Note  2.     Frequencies  shall  be  expressed  in  kilocycles  per  second  (kc/s)  at  and  below  30,000  kilo- 
cycles per  second  and  in  megacycles  per  second  (Mc/s)  above  this  frequency. 

Note  3.     Used  as  an  adjective  the  word  "Range"  shall  precede  the  number;  thus:  "Range  3." 


SECTION  2 
PROPERTIES  OF  MATERIALS 


CONDUCTING  MATERIALS 
ABTt  BY  KNOX  MCILWAIN  PAGE 

1.  Definitions 02 

2.  Properties  of  Specific  Conductors 03 

3.  Wire  Tables 12 

INSULATING  MATERIALS 
By  W.  R.  DOHAN 

4.  Dielectric  Properties 21 

5.  Solid  Dielectric  Materials 25 

6.  Liquid  Dielectrics 48 

7.  Gases  as  Dielectrics 53 


MAGNETIC  MATERIALS 

BY  TL  M.  BOZORTH  AND 

R.  A.  CHEGWIDDEN 

ART.  PAGE 

8.  Magnetic  Characteristics 57 

9.  High-permeability  Materials 60 

10.  Permanent-magnet  Materials 65 

11.  Magnetization  Curve 68 

12.  Effect  of  Temperature 69 

13.  Stress  and  Magnetostriction 70 

14.  Effect  of  Frequency 70 

15.  Measurement  of  Magnetic  Characteris- 

tics   72 


2-01 


PROPERTIES  OF  MATERIALS 
CONDUCTING  MATERIALS 

By  Knox  Mcllwain 

Since  all  materials  possess,  to  some  extent,  the  ability  to  conduct  electricity,  whether 
a  particular  material  is  called  conducting  or  insulating  is  a  matter  only  of  _  relative  degree. 
In  general,  if  a  moderate  potential  difference,  say  from  a  voltaic  cell,  is  placed  across 
a  section  of  a  material  and  a  measurable  current  flows  the  substance  is  said  to  be  con- 
ducting, but  if  no  conveniently  detectable  current  flows  it  is  considered  insulating. 

Among  conducting  materials  are  included  pure  metals,  some  metallic  salts  and  oxides, 
alloys,  and  the  metalloids,  carbon,  silicon,  and  boron.  Such  substances  as  glass,  dry  paper 
and  silk,  porcelain,  and  rubber  possess  such  a  very  low  conductivity  that  they  are  con- 
sidered insulating  materials. 

1.  DEFINITIONS 

Conductivity.  The  conductivity  of  a  material  is  the  direct  current  conductance  be- 
tween the  opposite,  parallel  faces  of  a  portion  of  the  material  having  unit  length  and  un.it 
cross-section. 

Effective  Conductivity.  The  effective  conductivity  of  a  material  to  a  periodic  current 
is  the  effective  conductance  between  the  opposite,  parallel  faces  of  a  portion  of  the  mate- 
rial having  unit  length  and  unit  cross-section. 

Resistivity.     The  resistivity  of  a  material  is  the  reciprocal  of  its  conductivity. 

Units  of  Resistivity.  The  resistivity  may  be  expressed  as  the  resistance  of  a  1-cm 
cube  of  material;  this  unit  is  called  the  ohm  per  centimeter  cube  or  preferably  simply  the 
ohm-centimeter.  In  the  English  system  the  ohm-inch  is  used.  When  the  material  is  to 
be  drawn  into  wires  the  ohm  per  mil-foot  is  used;  this  is  the  resistance  of  a  wire  1  mil 
(0.001  in.)  in  diameter  and  1  ft  long. 

Units  of  Conductivity.  The  conductivity  of  a  material  7  is  numerically  equal  to  the 
reciprocal  of  its  resistivity  p  and  is  expressed  in  mhos  and  megmhos  per  centimeter,  etc., 
instead  of  ohm-centimeters  and  microhm-centimeters,  etc.,  as  used  to  express  resistivity. 
Thus  y  equals  1/p. 

AnnealecL  Copper  Standard.  The  standard,  or  100  per  cent,  conductivity  is  defined 
'as  follows: 

1.  At  a  temperature  of  20  deg  cent,  the  resistance  of  a  wire  of  standard  annealed  copper 
1  meter  in  length  and  of  a  uniform  section  of  1  sq  mm  is  1/53  ohm  =  0.01724  ohm. 

2.  At  a  temperature  of  20  deg  cent,  the  density  of  standard  annealed  copper  is  8.89 
grams  per  cubic  centimeter.     This  corresponds  to  8.90  grams  per  cubic  centimeter  at 
0  deg  cent. 

3.  Thus,  at  20  deg  cent,  the  resistance  of  a  wire  of  standard  annealed  copper  of  uniform 
section,  1  meter  in  length  and  weighing  1  gram,  is  (1/55)  X  8-89  =  0.15328  ohm. 

Temperature  Coefficient  of  Electric  Resistance.  The  resistance  temperature  coeffi- 
cient j8<  of  a  substance  at  any  temperature  t  is  denned  as  the  rate  of  change  of  resistance 
at  this  temperature  divided  by  the  resistance  Rt  at  this  temperature: 


The  "mean"  temperature  coefficient  at  between  any  two  temperatures  t  and  ti  "referred 
to"  the  temperature  *  is  defined  as  the  "average"  change  in  the  resistance  in  this  interval 
per  degree  change  of  temperature,  divided  by  the  resistance  at  the  lower  temperature: 

=   Rg.  -  Rt 

The  temperature  coefficients  of  100  per  cent  conductivity  copper  are  given  in  Table  1 

2-02 


PROPERTIES  OF   SPECIFIC   CONDUCTORS 


2-03 


Table  1.    Temperature  Coefficients  of  Copper 


Ohms  per 
Meter-gram 
at  20°  C 

Per  Cent 
Conduc- 
tivity 

ao 

ai5 

CK20 

0:25 

«30 

0.  16134 
.  15966 
.15802 
.15753 
.15640 
.15482 
.15328 

95 
96 
97 
97.3 
98 
99 
100 

0.00403 
.00408 
.00413 
.00414 
.00417 
.00422 
.00427 
.00431 

0.00380 
.00385 
.00389 
.00390 
.00393 
.00397 
.00401 
.00405 

0.00373 
.00377 
.00381 
.00382 
.00385 
.00389 
.00393 

0.00367 
.00370 
,00374 
.00375 
.00378 
.00382 
.00385 
.00389 

0.00360 
.00364 
.00367 
.00368 
.00371 
.00374 
.00378 
.00382 

.  15176 

101 

.00397 

The  underlined  values  in  the  table  have  been  adopted  as  standard  by  the  American  Institute  of 
Electrical  Engineers. 


2.  PROPERTIES  OF  SPECIFIC  CONDUCTORS 

Conducting  materials  can  be  roughly  divided  into  two  groups:  the  good  conductors, 
and  the  resistive  conductors.  The  good  conductors  are  all  metals,  a  group  of  some  50 
chemical  elements  recognized  as  such  by  their  hardness,  ductility,  malleability,  luster, 
and  good  conductivity  of  heat  and  electricity.  Since  these  and  other  characteristics  are 
possessed  by  these  elements  in  varying  degrees  there  are  some  which  are  metallic  in 
some  properties  and  non-metallic  in  others.  Of  these  an  electrically  important  group  is 
carbon,  silicon,  and  boron,  often  called  the  metalloids,  which  are  rather  good  conductors 
of  heat  and  electricity  but  have  non-metallic  mechanical  properties.  Almost  all  the 
resistive  conductors  on  the  market  today  are  solid  solution  alloys  or  those  composed 
largely  of  solid  solutions.  In  electric  circuits  they  are  generally  used  either  in  devices 
for  purposes  of  operation,  protection,  or  control,  or  as  heating  elements.  It  is  usually 
desirable  for  them  to  have  properties  of  high  resistivity  and  low  temperature  coefficient 
of  resistance,  but  in  some  cases  a  high  temperature  coefficient  is  useful — witness  the  use 
of  nickel  as  a  filament  control. 

In  general,  the  standard  alloys  for  electrical  resistance  are  made  of  nickel  and  chromium, 
compositions  of  80  per  cent  nickel  and  20  per  cent  chromium  having  high  resistance  to 
oxidation  with  maximum  working  temperatures  up  to  about  1100  deg  cent.  By  varying 
the  proportions  of  these  and  by  the  addition  of  different  amounts  of  iron,  copper,  man- 
ganese, zinc,  and  cobalt  we  are  able  to  obtain  alloys  which  have  different  resistivities, 
temperature  coefficients,  melting  points,  magnetic  properties,  and  heat-  and  corrosion- 
resisting  properties. 

Tables  2  and  3  give  the  properties  of  materials  available  for  the  manufacture  of  con- 
ductors and  resistors;  Table  4  gives  the  physical  properties  of  some  beryllium-copper  alloys. 


2-04 


PROPERTIES  OF  MATERIALS 


Table  2.    Properties  of  Conducting  Materials  at  Usual  Temperatures 


Description  of  Material 

Resistivity 
Microhm-cm 

Temperature 
Coefficient, 
See  Note  a 

Max. 
Work- 
ing 
Temp., 

°C 

Den- 
sity, 
jm 

Tensile 
Str. 
(an- 
nealed), 
Ib 
in.2 

Coef. 
Lin. 
Expan- 
sion 
per  °C 
10~6X 

0°C 

20°  C 

Temp., 
°C 

a 

Acme  *  (Ni  30  +  Cr  5  +  Fe  65) 
Advance  *  (Ni  45  +  Cu  55)... 

Alferon*(Cr  14.25  +  A1  3.5  + 
Fe8225)  

(Similar  Chr 
(Similar  Chr 
(Similar 

33.3 
2.62 

2.607 

8.85 
39.1 

28.5 
35 

3.64 
6.29 

17.8 
19.0 

(800- 
1,300) 
4,000 

2.6 

(Similar  Chr 
(Similar 

87.3 
48.8 
83.1 

112 

omel  A) 
omel  C) 
Comet) 

2.828 
(15°)  42 

10.1 
120 

8  X  1012 

7.60 

4.59 
110 

100 
108 
112 
99.5 

omel  A) 
-Phenix) 

20 
20-100 
20 

20-500 

0 
0-100 

18 

15 
20 

0-160 

(Simila 
Sil 

0-100 
20 

0-100 
0-100 

19-92 
20 

20 

25-387 
25-335 
20-500 
20-500 
20-500 
20-500 

0.00072 
.00002 
.0006 

.00016 

.0012 
.00423 

.0039 

.000897 
.0036 

.000387 

r  German 
ver) 
.0042 

.00424 
.004 

.00204 
.00158 

.0005 
.0038 

.0036 
.00025 

(-.0006 
-.0012) 
-.0003 

.00031 
.00013 
.00017 
.00032 

1,000 
535 
600 

1,100 

1,250 
300 

1,230 

1,000 

400 
350 
5,100 
1,100 
900 
500 

8.15 
8.9 

7.31 
2.7 

8.15 

1.56 
1.54 
7.95 
8.4 
8.24 
7.94 

100,000 
60,000 
80,000 

100,000 
35,000 

90,000 

500 
600 
70,000 
95,000 
95,000 
70,000 

14.9 
14 

11.4 
24 

16 

15.8 
17 
17 
16 

Alloy  A       

Alloy  C  

Alloy  D                       

Alumel  *  (Ni  94  +  Mn  2.5  + 
Fe0.5  +  A12  +  Sil)  
Aluminum  (Pure) 

Aluminum  (Wire,  61%  cond.). 
Aluminum  Bronze  (Cu  97  + 
A13) 

Antimony        

Argentan*(Cu61.6  +  Nil5.8 
+  Zn  22  6) 

Argentan*(Cu56  +  Ni26  + 
Znl8) 
Arsenic  

Ascoloy*(Fe82-S6  +  Crl6- 
05)        

Beryllium  

Bismuth 

Boron 

Brass  (Cu  90.9  +  Zn  9.1)  
Brass  (Cu  65.8  +  Zn  342)  .... 

Bronze  (Cu  88  +  Sn  12) 

Cesium  

Calcium  (9937%  pure)  

Calido*(Ni59  +  Crl6  +  Fe 
25)     

Calorite  *  (Ni  65  +  Cr  12  + 
Fe  15  +  Mn8). 

Carbon  (graphite) 

Carbon  (incandescent  lamp)  .  . 
Chromax  *  (Ni  30  +  Cr  20  + 
Fe  50)  . 

Chromel  *  A  (Ni  80  +  Cr  20) 
Chromel  *C  (Ni  60  +  Cr  16  + 
Fe24) 

Chromel  *D(Ni  30  +  Cr  20 
+  Fe50)  

Chronin  *  (Ni  83.7  +  Cr  14.7} 
Cimet  *  (Ni  25  +  Fe  75)  .  .  .  . 

range  is  given,  the  coefficient  a  is  the  mean  value  for  the  range,  referred  to  the  lower 


*  Trademark  names. 


PROPERTIES  OF  SPECIFIC  CONDUCTORS 


2-05 


Table  2.    Properties  of  Conducting  Materials  at  Usual  Temperatures — Continued 


Description  of  Material 

Resistivity 
Microhm-cm 

Temperature 
Coefficient, 
See  Note  a 

Max. 
Work- 
ing 
Temp., 
°C 

Den- 
sity, 

Tensile 
Str. 
(an- 
nealed), 
Ib 
in.2 

Coef. 
Liu. 
Expan- 
sion 
per  °C 
10~6X 

0°C 

20°  C 

Temp., 
°C 

a. 

Climax*  (Ni  25  +  Fe  74  + 
Mn  1)         

(Similar 
49.0 

1.589 
1.56 
1.60 

4.08 
100 
77 

48 

(Similar 
(Similar 

(Similar 
(Similar 

47 

91.4 

27.1 
53 

33.1 
89,000 
2.22 

87 
9.7 

95 

Advance) 
49 

48.8 
1.7241 

1.77 

Lucero) 
Comet) 

Advance) 

Invar) 
83.2 

8 
133 

92 

49.2 
28.2 

33.8 

(Similar 

20 
0-100 
0 

20-500 

0-100 
12 
25 
100 
200 
500 
0-100 

0-100 
20 
0-100 
100 
0-100 
20 

0-100 
0-100 
0 

0 

20 

0 
-50  to 
100 

20 

20 
20 

20 

18-100 
20 

Chromel 

.0007 
.00658 
.0033 

.00088 

O.OOOOi 
.000008 
.000002 
-.000033 
-.000020 
.000027 
.00002 

.00427 
.00393 
.00428 
.0038 
.00408 
.00382 

.00155 
.00004 
.000022 

-.00003 

.0011 

.00005 
±0.00002 

.00016 

.0000 
.00207 

.00031 

.00368 
.0034 

C) 

600 

500 
500 

600 

900 
1,150 
340 

260 
500 

8.15 

8.86 
8.92 

8.1 

8.9 
7.8 

8.5 
19.3 

55,000 
62,000 

60,000 
35,000 

45,000 

100,000 
20,000 

15 

14.9 
17 

16.6 

14 

17.3 
14.2 

Cobalt  (99  8%  pure) 

Comet  *  (Ni  30  +  Cr  4.75  + 
Fe  65  25) 

Constaloy*(Ni45  +  Cu55).. 
Constantan  *  (Cu  60  +  Ni  40) 

Copel  *  (Ni  45  +  Cu  55)  
Copper  (annealed  standard)  ,  .  . 
Copper  (electrolytic)             .  . 

Copper  (hard-drawn)    

Copper-iron  (Fe  0  4%) 

Copper-manganese  (Cu  70  + 
Mn  30)                         

Copper-manganese-iron      (Cu 
70.6  +  Mn  23.2  +  Fe  6.2).. 

Copper-manganese-nickel  (Cu 
73  +  Mn  24  +  Ni  3)  
Corronil  *  (Ni  70  +  Cu  26  + 
Mn4)              

Cronin  *  D  

Cronit  *  (Ni  60  +  Cr  40)  

Dilver  *  (similar  Invar) 

Dumet*(Ni46  +  Fe54).... 
Electris  *  (similar  Phenix)  

Elinvar*  (Ni  36  +  Cr  12  + 
Fe  52)                         .... 

Eureka  *  (similar  Lucero)  
Evanohm  *  (Cr  20  +  Al  2.5  + 
Cu  2  5  +  Ni  bal  ) 

Excello*  (Ni  85  +  Cr  14  + 
Fe0.5  +  Mn0.5)  

Excelsior  *  (similar  Advance)  .  . 
Ferro-nickel  

Gallium 

German  silver  fl  8%  (Ni  18  + 
Cu64  +  Zn  18) 

GerionfvniiiTfl          ,     .  »   .  ,    ,  , 

Gold  (99.9%)  

Glowray*  (Ni  65  +  Cr  12  + 
Fe23)  

NOTE  a:  Where  a  temperature  range  is  given,  the  coefficient  a  is  the  mean  value  for  the  range,  referred  to  the  lower 
temperature. 

*  Trademark  names. 

t  German  Silver-30%  has  substantially  the  same  properties  as  "Advance." 


2-06 


PROPERTIES  OF  MATERIALS 


Table  2.    Properties  of  Conducting  Materials  at  Usual  Temperatures— Continued 


Description  of  Material 

Resistivity 
Microhm-cm 

Temperature 
Coefficient, 
See  Note  a 

Max. 
Work- 
ing 
Temp., 
°C 

Den- 
sity, 

cm3 

Tensile 
Str. 
(an- 
nealed), 

A 
in.2 

Coef. 
Lin. 
Expan- 
sion 
per  °C 

io-flx 

0°C 

20°  C 

Temp., 
°C 

a. 

Hipernik  *  (Ni  50  4-  Fe  50)  ... 
Hopkinson  alloy  *  (Ni  25  +  Fe 
75)     

47.1 
50.2 

49 

(Similar 
8.37 

(Similar- 

6.10 
8.85 

11.8 
45.7 

74,4 
97.8 

85.0 
19.8 

63.3 
8.55 

4.35 
5.0± 

100 

(Similar 

20 
47.1 

50.2 

91.61 

Nirex) 

Platinite) 
10 

135 

103 

94.6 
22 

10 
48.2 
4.6 

14 
48.2 

20 

Phenix) 

20-100 
20 

20 

0-100 
20 

0 

0-100 
0-100 
100 
20 

10-35 
10-35 
0 

20-500 
20 

20 

0-100 
20 

0 
20-100 

20-250 
20 
0 

0-100 

20-100 

15-35 
25 
100' 
20-100 

,0045 
.000005 

.000011 

.0000± 
.000005 
.000479 

0.0047 

.00411 
.00625 
.0068 
.0050 

.00423 
.00161 
.0016 

.00003 
.00016 

.000242 
.00070 
.00411 
.0039 

,0047 
.00071 

.0010 
.004 
.0038 

.00004 

.0045 

.000015 
.000000 
—.000042 
.0036 

500 

520 
150 

1,000 

900 
1,100 

1,100 

700 
600 

1,100 
100 

1,100 

8.46 
8.92 

8.92 
8.9 

7.86 

8.9 

8.9 
8.9 

8.8 
8.2 

8.75 

70,000 
65,000 

80,000 

100,000 
90,,  000 

80,000 

70,000 
100,000 

90,000 
60,000 

90,000 

15 

14 

1.5 

As  glass 

11.7 

16 
15 

18 
12.5 

14.6 
18.7 

14.3 

Hytemco*(Ni724-Fe28).. 
la  la*  (Ni40H-Cu60soft).. 
la  la  *  (Ni  40  4-  Cu  60  hard- 
drawn)                        •     •  •  • 

Ideal  *  (Ni  40  4-  Cu  58  4-  Fe 
1  4-  Mn  1) 

Inconel  *  (Or  13  4-  Fe  8  4-  Ni 
49)  

Indium           

Invar  *  (Ni  36  4-  Fe  64) 

Invariant  *  (Ni  47  4-  Fe  53)  .  . 
Iridium                    

Iron  (pure)  

Iron  (99.98%  pure)  

Iron  (steel)  soft        ,          ... 

Iron  (steel)  tempered  glass  hard 
Iron,  cast  (soft) 

Iron  cast  (hard) 

KanthalD(Cr234-A134- 
Co2  4-  Fe  bal.)       

Karma  *  (Ni  80  4-  Cr  20)  .  .  .  . 
Kromax  *  (similar  Karma)  .... 

Kromore  *  (Ni  85  +  Or  15)  .  .  . 
Krupp  metal  (nickel  steel)  
Lead  

Lead-bismuth  (Pb  42.3  4-  Bi 
57,7)  

Lohm  *  (Ni  6  4-  Cu  94) 

Lucero  *  (Ni  70  4-  Cu  30)  .  .  .  . 
Magno  *  (Ni  95  4-  Mn  5)  
Magnesium 

Magnesium  (free  from  zinc)  .  .  . 

Manganese-copper  *  (Cu  70  4- 
Mn30)  

Manganese-nickel  (Mn  2  4-  Ni 
98)  

Manganin*  (Cu84-f-Mn  12 
4-N14)  

Magno*(Mn4.54-Ni95.5).. 

temperItoeWhere  *  ^^^^  range  is  given' the  coefficieilt  «  is  **  *ean  value  for  the  range,  referred  to  the  lower 
""Trademark  names. 


PROPERTIES  OF  SPECIFIC  CONDUCTORS 


2-07 


Table  2.    Properties  of  Conducting  Materials  at  Usual  Temperatures — Continued 


Description  of  Material 

Resistivity 
Microhm-cm 

Temperature 
Coefficient, 
See  Note  a 

Max. 
Work- 
ing 
Temp., 
°C 

Den- 
sity, 
gm 
cm3 

Tensile 
Str. 
(an- 
nealed), 
Ib 
m7 

Coef. 
Lin. 
Expan- 
sion 
per°C 
10~6X 

0°C 

20°  C 

Temp., 
°C 

a 

Marsh's  patent  *  (Ni  75  +  Or 
25)  

(Similar  Chr 
94.07 

5.14 

4.2 
4.9 

(Similar 
40.8 

6.93 
9.9 

29.4 

10.21.. 

7.75 

(Similar 

45 
9.83 

omel  A) 
95.783 

30 

Lucero) 
42.6 

112 
108 

7.8 

7.236 

33.3 
48.2 

80.5 

98.1 
166 
9.5 

11 

95.5 
83.1 

Premier) 

34.4 
(18°) 

0-100 
20 
20-100 

0-100 
25 
0-170 
0-170 

20 

20-500 
20-500 
20 
0-100 

20 

20 
20 

20-100 

20-500 
20-500 

20 
0 

20 

20 
0 

0 
20 

Noteb 
.00089 
.00018 

.00435 
.0033 
.0050 
.0050 

.0020 

.00017 
0.00013 
.006 
.00618 

.00400 

.00027 
.00020 

.00135 

.00012 
.000066 

.0033 
.0035 

.00018 

.0011 
.0040- 

.003 
.003 

700 

425 

980 
1,100 

700 

260 
260 

500 

1,100 
500 

1,100 
400 

8.9 

8.15 

8.25 
8.41 

8.8 

8.5 
8.5 

8.08 

8.55 
6.8 

8,05 
8.10 

70,000 

44,000 

120,000 
95,000 

70,000 

60,000 
60,000 

100,000 

130,000 
130,000 

75,000 

17.5 

12.6 

17 
17 

14,0 
17.3 

1.0 

16.1 
15.8 

14 

Mercury  

Midohm  *  (Ni  23  +  Cu  77)  .. 
Molybdenum  (very  pure)  

Molybdenum  (annealed)  
Molybdenum  (hard-drawn)  .  .  . 

Mond  *  No.  70  (Ni  70  +  Cu 
26  +  Mn4)  

Monel  metal  (Ni  67  +  Cu  28 
+  Mn5) 

Nichrome*(Ni61  +  CH5  + 
Fe24)  

Nichrome  *  V  (Ni  80  +  Cr  20) 
Nickel  

Nickel  (electrolytic)  

Nickel  (very  pure)  

Nickel  (commercial  wire)  
Nickel-chromium  (Ni  +  Cr 
and  Fe  +  Mu)  

Nickel-silver  18%  (Ni  18  + 
Cu64  +  Zn  18)        

Nickel-silver  30%   (Ni  30  + 
Cu  50  +  Zn  20)  

Nickel  steel  (4.35%  Ni)  

Nickelin  *  (same  German  sil- 
ver) 
Nicraloy  *  (similar  Chromel)  .  . 
Nilvar  *  (Ni  36  +  Fe  64)  

Nirex*  (Cr  13  +  Fe  8  +  Ni 
79)       

Ohmax  *      

Osmium  

Palladium  

Palladium  (very  pure)  

Peerless  *  (Ni  78.5  +  Cr  16  + 
Fe  3  +  Mn  2) 

Phenix  *  (Ni  25  +  Fe  75)  
Phosphor-bronze  

Placet  *  (Ni  60  +  Cr  15  +  Fe 
20  +  Mn5)  

Platinite  *  (Ni  42-46,  Fe  58- 
54)  

Platinoid  *  (Cu  62  +  Ni  15  + 
Zn  22) 

Platinum  

NOTE  a:  Where  a  temperature  range  is  given,  the  coefficient  a  is  the  mean  value  for  the  range,  referred  to  the  lower 
temperature. 

NOTE  b:  Use  equation  Rt  =  RQ  (I  +  aT  +  6T2)  with  a  =  0.0008649  and  6  =  0.00000112. 
*  Trademark  names. 


2-08 


PROPERTIES  OF  MATERIALS 


Table  2.    Properties  of  Conducting  Materials  at  Usual  Temperatures— Continued 


Description  of  Material 

Resistivity 
Microhm-cm 

Temperature 
Coefficient, 
See  Note  a 

Max. 
Work- 
ing 
Temp., 
°C 

Den- 
sity, 
gm 

Tensile 
Str. 
(an- 
nealed), 
Ib 
in.2 

Coef. 
Lin. 
Expan- 
sion 
per  °C 
10~6X 

0°C 

20°  C 

Temp., 
°C 

a 

cm3 

Platinum,  drawn,  wire  

10.96 
31.6 

21.14 
6.1 

(Similar 
44.6 

53 
5.11 

64.5 
11.6 

1.468 
4.3 

14.6 

17.6 

46.7 

(Similar 
10.5 

(Similar  Chr 
(Similar  Chr 
(Similar  Chr 

103.0 
133 

95.7 
Rayo) 

25 
113.0 

58=b 
1.629 
(18°) 

73 

24.8 

103.0 
15.5 

41 
.2  X  I06 
(19.68) 

46.7 
Phenix) 
11.5 

melA) 
melC) 
melD) 

5.51 

0-100 
0 
-100  to 
+  100 
0 

15 

0 
20 
20-500 
20 

0 

0 
0 

0-94.3 
0 

20-250 
20 

20 
0-100 
0 
20-500 

20 
0-100 
20 

20 

20 

0-100 
20 

18 

.00367 
.0037 

.002± 
.0008 

.00143 
.0057 
.00036 
.0001 
.00018 

.00041 

0.0004 
.0043 

.0023 
.0060 

.0027 
.000025 

.0038 
.00400 
.0054 
.00094 

.00011 
.0033 
.0031 

.000025 

56X10~7 

.00465 
.0042 

.0045 

1,200 

1,000 
500 
1,100 

1,100 
1,100 

500 

1,100 

1,100 
100 

200 
2,000 

21.45 

8.15 
7.30 
8.05 

8.72 
7.63 

10.5 

7.93 

8.2 
8.89 

8.15 
19.3 

50,000 

90,000 
90,000 

130,000 
42,000 

200,000 

78,000 
600,000 

8.9 

15.5 
15 

15.2 
18.9 

20 

19.4 
4 

Platinum-indium  (Pt  80  +  Ir 
20) 

Platinum-rhodium   (Pt  90  + 
Rh  10) 

Potassium     

Premier  *  (Ni  61  +  Cr  11  + 
Fe  25  +  Mn  3)    

Radiohm*(Crl6.5  +  AI5  + 
Fe  78  5) 

Rayo*(Ni85+Cr15)  
Redray*(Ni85  +  Crl5).... 
Rheotan  *  (Ou  84  +  Fe  12  + 
Zn  4)                       

Rheotan  *  II  (Cu  53.3  +  Ni 
25.3  +  Fe4.5  +  Znl6.9)... 
Rhodium 

Rose's  metal  *  (Bi  48.9  +  Sn 
233  +  Pb  27.6)  

Rubidium 

R-63  Alloy  (Mn  4  +  Si  1  + 
Ni  95)  

Silchrome  *  (Si  +  Cr  +  Fe)  .  .  . 
Silicon  .  .         

Silver  (99  78%  pure) 

Silver  (electrolytic)   

Sodium 

Stainless  Type  304  <Cr  18  + 
Ni  8  -f  Fe  74) 

Steel  (see  iron) 

Rfcrnntiufn  .  T  -.--,-    ,  ,    .  .    -  ,  r 

Superior  *  (Ni  78  +  Cr  19.5  + 
Fe  0.5  +  Mn  2) 

Tantalum     

Tarnac  *  (similar  Manganin)  .  . 
Tellurium  ,  t  ,.,.,. 

Thallium  (pure) 

Therlo  *  (Cu  +  Mn  +  Al)  .  .  .  . 
Ticjo*(Ni  27.5  -hFe  72.5).... 
Tin  

Tophet*A  

Tophet*C  

Tophet  *  D  

Tungsten  

NOTE  a:  Where  a  temperature 
temperature* 

*  Trademark  names. 


range  is  glven,  the  coefficient  a  is  the  mean  value  for  the  range,  referred  to  the  lower 


PROPERTIES   OF   SPECIFIC   CONDUCTORS 


2-09 


Table  2.     Properties  of  Conducting  Materials  at  Usual  Temperatures — Continued 


Description  of  Material 

Resistivity 
Microhm-cm 

Temperature 
Coefficient, 
See  Note  a 

Max. 
Work- 
ing 
Temp., 
°C 

Den- 
sity, 
gm 
cm3 

Tensile 
Str. 
'    (an- 
nealed), 
Ib 
ia.2 

Coef. 
Lin. 
Expan- 
sion 
per°C 
10~6X 

0°C 

20°  C 

Temp., 
°C 

a 

Tungsten  (annealed) 

51.8 

5.38 
5.75 

(Similar 

(Similar 
(Similar 

(Similar  Eva 

4.37 

33.0 
5.92 

93.1 
5.0 
66.5 

Advance) 
45.7 
43.2 
Lohm) 
15 
90  Alloy) 

8 

87.2 
nohm) 
100 

0-170 
0-69.8 

20 
18-100 
20 

20-500 
20-100 
20-500 

20-500 
20-500 

20-100 

0-100 
20 
20-500 

.0051 
.0023 

.000155 
.00402 
.00347 

.0025 
.0013 
.0012 

.0027 
.0029 

.00049 

.0060 
.00072 
.00034 

100 

1,100 
500 
1,100 

1,100 
1,100 

500 

650 
500 

8.6 
7.14 

8.10 
8.9 
8.12 

8.17 
8.25 

8.9 
8.15 

25,000 

100,000 
30.000 
100,000 

100,000 
100,000 

35,000 
60,000 

15.9 
33 

10.8 
17.5 
5.3 

8.0 
9.5 

17.5 

17.1 
13.1 

Wood's  metal  *  (Bi  55.7  +  Sn 
13.7  +Pbl  3.7  +  Cd  16.2). 

Yankee  silver  *  (similar  Nickel 
silver)                       

Zinc  (pure) 

Zinc  (trace  Fe) 

14  Alloy  (Ni  42  +  Or  5.5  + 
Fe52.5)  

30  Alloy  (Ni  2.25  +  Cu  97.5). 
42  Alloy  (Ni  42  +  Fe  58)  .... 

45  Alloy 

46  Alloy  (Ni  46  -h  Fe  54)  
52  Alloy  (Ni  51  +  Fe  49)  
60  Alloy 

90  Alloy  (Ni  1  1  +  Cu  89)  .  .  .  . 
95  Alloy                

99  Alloy  (Ni  99.8)  

193  Alloy  (Ni  30  +  Cr  2  + 
Fe  67  +  Mn  1  ) 

331  Alloy  

525  Alloy  

NOTE  a:  Where  a  temperature  range  is  given,  the  coefficient  a  is  the  mean  value  for  the  range,  referred  to  the  lower 
temperature. 
*  Trademark  names. 


2-10  PROPERTIES  OF  MATERIALS 

Table  3.    Properties  of  Conducting  Materials  at  High  Temperatures 


Description  of  Material 

Resistivity  in  Microhm  centimeters 

500°  C 

1000°  C 

1500°  C 

Al      *          (f      d) 

10 

152 
139.9 

60X106  appx. 
12.5 

34.12 
109 
2700 

3700 
3300 
2800 

8.  5X1  OB  appx. 
4.8X106  appx. 
0.22X106 

5.1 
2.50X106 

5640X106 
1570X106 

94 
330  XI  06  appx. 

6.62 
840 
800 

2.70X106 
52  appx. 

1260X106 
115 
10X106 

102.85 
0.418X106 
0.824X106 

81 

2200X106 
16.5 

t 

119 
34.4 
25.3 

19.7X106 

24 
8X109 
136 
167.5 

41 

122 
2400 

3400 
3000 
2100 

2.8X106 
1.9X106 
0.12X106 

9.42 

18X106 

105 
IX  106  appx, 

12.54 
860 
650 

1.7X106 
1  1  1  appx. 

31.4X106 
4.8X106 
125 

98 
1400X106 
15.7X106 

28.5 

128 
66 
40.8 

15X106  appx. 
110X106 
3.7X106 

29 
75X107 

136 
2200 

2900 
1600 

0.85X106 
24.8 

37 

890 
580 

1.2X106 
131  appx. 
166 

3.4X106 
148 

40.5 
0.5X106  appx. 

98 
52.6 

o  5  vine 

Brass  (2   1  fused)  

Brass  (2   I  solid)                   

Calido  (solid)                         

Carbon  (a)            

Carbon  (b)                            

Copper  (solid}                         . 

Copper  chloride  (fused)  .  

Copper  oxide  (Cu  0)           

Copper  oxide  (Cu  O«>  powder),.  ,,.  .... 

Ferro-nickel  (solid)  .        

Glass      

Gold  (fused) 

Gold  (solid)  

Graphite  (a)  ... 

Graphite  (b)  

Iron  (a)  solid  .                       

Iron  (b)  fused  

Iron  oxide  (Fe^>  Og,  powder)  

Krupp  metal  (solid)  

Lead  (fused)  

Lead  chloride  (fused  520°)  

Lead  chloride  (solid)  

Lead-tin  alloy  (fused)  

Magnesium  oxide  (powder)  

Manganese  oxide  (powder)  

Manganese  oxide  (Mn  C>2»  powder)  

Molybdenum  (solid)  

KTArnst.  filq.Tnpnf,  IL 

Nichrome*  II  (solid)  

Platinum  (a)  solid  

Platinum  (b)  solid  

Porcelain  

Quartz  

Refrax*  

*  Trademark  names. 

PROPERTIES  OF  SPECIFIC   CONDUCTORS 


2-11 


Table  3.    Properties  of  Conducting  Materials  at  High  Temperatures — Continued 


Description  of  Material 

Resistivity  in  Microhm-centimeters 

500°  C 

1000°  C 

1500°  C 

Silfrax*  B  

0.92X106 
0.094X106 
to   0.  023X106 
120X106 

5 
0.547X106 

36 

54.62 

18 
18 
36.60 

0.84X106 

3.5X106 
17.01 

0.90X106 
57 

68 

30.5 
33.4 

26.7X106 

0.7X106 
23 

78 

74 
80.5 
43 

50 

Silver  (fused)      

Silver  (solid)  

Silver  chloride  (fused)  .    .. 

Sodium  chloride  (fused)  ...            .  .              .    » 

Tantalum  (solid)  

Tantalum  (a)  solid                                              ,    . 

Tantalum  (b)  solid  

Tin  (fused) 

Tungsten  (solid) 

Tungsten  (a)  solid  

Tungsten  (b)  solid 

Zinc  oxide  (powder)  

*  Trademark  names. 


Table  4. 


Approximate  Values  for  the  Physical  Properties  of  Beryllium-copper  Alloys 
of  the  21  Per  Cent  Beryllium  Class 


Condition 

Solution- 
treated 
"Annealed" 

Solution- 
treated  and 
Cold-worked 

Solution-treated 
and 
Precipitation- 
hardened 

Solution-treated, 
Cold-worked,  and 
Precipitation- 
hardened 

Electrical  conductivity  %  I.A.- 
C.S.  at  20°  C 

17 

17 

(a)     20-25 

(a)     20-25 

Tensile  strength,  psi        

70  ,  000 

90,000 

(&)      32-38 
(a)    160  000 

(6)      32-38 
(a)    180  000 

Yield    strength,    psi,    at    0.5% 
elongation  under  load      .  . 

(6)    130,000 
(a)    150  000 

(6)    140,000 
(a)    175  000 

Elongation  in  2  in  ,  % 

35 

(a)       3  0 

(a)       2  0 

Modulus  of  elasticity,  psi  .  .    . 

16  X  106 

18  4-19  4  X  106 

Endurance  limit,  psi,  at  108  re- 
versals of  stress  

23,000 

28,000 

28,000 

(a)  Heat-treated  for  maximum  hardness.    (6)  Heat-treated  for  maximum  conductivity. 


2-12 


PROPERTIES  OF  MATERIALS 


3.  WIRE  TABLES 

Table  5.    Solid  Copper  Wire 

A.  W.  G.  or  B.  &  S.  Gage;  English  Units 
100  per  cent  conductivity;  density  8.89  at  20  deg  cent 


Gage 
No. 

Diam- 
eter in 
Mils 

Cross-section 

Resistance  at 
20°  C  or  68°  F 

Weight  in  Pounds 

Feet 
per 
Pound 

Circular 
Mils 

Square 
Inches 

Ohms  per 
1000ft 

Ohms 
per  Mile 

per 
1000ft 

per 
Mile 

0000 

460.0 

211,600 

0.1662 

0.0490 

0.25 

640.5 

3380 

1.561 

000 

409.6 

167,800 

0.  1318 

0.0618 

0.32 

507.9 

2680 

1.968 

00 

364.8 

133,100 

0.1045 

0.0779 

0.41 

402.8 

2130 

2.482 

0 

324.9 

105,500 

0.08289 

0.9082 

0.51 

319.5 

1680 

3.130 

1 

289.3 

83,690 

0.06573 

0.1239 

0.65 

253.3 

1340 

3.947 

2 

257.6 

66,370 

0.05213 

0.1563 

0.82 

200.9 

1060 

4.977 

3 

229.4 

52,640 

0.04134 

0.1970 

1.04 

159.3 

841 

6.276 

4 

204.3 

41,740 

0.03278 

0.2485 

1.31 

126.4 

667 

7.914 

5 

181.9 

33,100 

0.02600 

0.3133 

1.65 

100.2 

529 

9.980 

6 

162.0 

26,250 

0.02062 

0.3951 

2.09 

79.46 

420 

12.58 

7 

144.3 

20,820 

0.01635 

0.4982 

2.63 

63.02 

333 

15.87 

8 

128.5 

16,510 

0.01297 

0.6282 

3.32 

49.98 

264 

20.01 

10 

101.9 

10,380 

0.008155 

0.9989 

5.28 

31.43 

166 

31.82 

12 

80.81 

6,530 

0.005129 

1.588 

8.38 

19.77 

104 

50.59 

14 

64.08 

4,107 

0.003225 

2.525 

13.3 

12.43 

63.3 

80.44 

15 

57.07 

3,257 

0.002558 

3.184 

16.8 

9.858 

52.0 

101.4 

16 

50.82 

2,583 

0.002028 

4.015 

21.2 

7.818 

41.3 

127.9 

17 

45.26 

2,048 

0.001609 

5.064 

26.7 

6.200 

32.7 

161.3 

18 

40.30 

1,624 

0.001276 

6.385 

33.7 

4.917 

26.0 

203.4 

19 

35.89 

1,288 

0.001012 

8.051 

42.5 

3.899 

20.6 

256.5 

20 

31.96 

1,022 

0.0008023 

10.15 

53.6 

3.092 

16.3 

323.4 

21 

28.46 

810.1 

0.0006363 

12.80 

67.6 

2.452 

12.9 

407.8 

22 

25.35 

642.4 

0.0005046 

16.14 

85.2 

1.945 

10.3 

514.2 

23 

22.57 

509.5 

0.0004002 

20.36 

108 

1.542 

8.14 

648.4 

24 

20.10 

404.0 

0.0003173 

25.67 

135 

1.223 

6.46 

817.7 

25 

17.90 

320.4 

0.0002517 

32.37 

171 

0.9699 

5.12 

1,031 

26 

15.94 

254.1 

0.0061996 

40.82 

216 

0.7692 

4.06 

1,300 

27 

14.20 

201.5 

0.0001583 

51.46 

272 

0.6100 

3.22 

1,639 

28 

12.64 

159.8 

0.0001255 

64.90 

343 

0.4837 

2.55 

2,067 

29 

11.26 

126.7 

0.00009953 

81.84 

432 

0.3836 

2.03 

2,607 

30 

10.03 

100.5 

0.00007894 

103.2 

545 

0.3042 

1.61 

3,287 

31 

8.928 

79.70 

0,00006260 

130.1 

687 

0.2413 

1.27 

4,145 

32 

7.950 

63.21 

0.00004964 

164.1 

866 

0.1913 

1.01 

5,227 

33 

7.080 

50.13 

0.00003937 

206.9 

1,090 

0.1517 

0.814 

6,591 

34 

6.305 

39.75 

0.00003122 

260.9 

1,380 

0.  1203 

0.635 

8,310 

35 

5.615 

31.52 

0.00002476 

329.0 

1,740 

0.09542 

0.504 

10,480 

36 

5.000 

25.00 

0.00001964 

414.8 

2,190 

0.07568 

0.400 

13,210 

37 

4.453 

19.83 

0.00001557 

523.1 

2,762 

0.06001 

0.317 

16,660 

38 

3.965 

15.72 

0.00001235 

659.6 

3,480 

0.04759 

0.251 

21,010 

39 

3.531 

12.47 

0.000009793 

831.8 

4,392 

0.03774 

0.199 

26,500 

40 

3.145 

9.888 

0.000007766 

049 

5,540 

0.02993 

0.158 

33,410 

41 

2.800 

7.842 

0.000006159 

323 

6,983 

0.02374 

0.125 

42,130 

42 

2.494 

6.219 

0.000004884 

668 

8,806 

0.01882 

0.0994 

53,120 

43 
44 

2.221 
1.978 

4.932 
3.911 

0.000003873 
0.000003072 

2103 
2652 

11,100 
14,000 

0.01493 
0.01184 

0..  0788 
0  0625 

66,990 
84,470 

WIRE   TABLES 


2-13 


Table  6.    Solid  Copper  Wire 

A.  W.  G.  or  B.  &  S.  Gage  in  Metric  Units 

100  per  cent  conductivity;  density  8.89  at  20  deg  cent 


Gage  No. 

Diameter, 
mm 

Cross-section, 
sq  mm 

Ohms  per  Kilometer 
20°  C 

Kilograms  per 
Kilometer 

0000 
000 
00 

11.68 
10.40 
9.266 

107.2 
85.03 
67.43 

0.1608 
0.2028 
0.2557 

953.2 
755.9 
599.5 

0 

2 

8.252 
7.348 
6.544 

53.48 
42.41 
33.63 

0.3224 
0.4066 
0.5126 

475.4 
377.0 
299.0 

3 

4 
5 

5.827 
5.  189 
4.621 

26.67 
21.15 
16.77 

0.6464 
0.8152 
1.028 

237.1 
188.0 
149.1 

6 
7 
8 

4.115 
3.665 
3.264 

13.30 
10.55 
8.366 

1.296 
1.634 
2.061 

118.2 
93.78 
74.37 

10 
12 
14 

2.588 
2.053 
1.628 

5.261 
3.309 
2.081 

3.277 
5.211 
8.285 

46.77 
29.42 
18.50 

15 
16 
17 

1.450 
1.291 
1.150 

1.650 
1.309 
1.038 

10.45 
13.18 
16.61 

14.67 
11.63 
9.226 

18 
19 
20 

1.024 
0.9116 
0.8118 

0.8231 
0.6527 
0.5176 

20.95 
26.42 
33.31 

7.317 
5.803 
4.602 

21 
22 
23 

0.7230 
0.6438 
0.5733 

0.4105 
0.3255 
0.2582 

42.00 
52.96 
66.79 

3.649 
2.894 
2.295 

24 
25 
26 

0.5106 
0.4547 
0.4049 

0.2047 
0.1624 
0.1288 

84.22 
106.2 
133.9 

1.820 
1.443 
1.145 

27 

28 
29 

0.3606 
0.3211 
0.2859 

0.1021 
0.08098 
0.06422 

168.8 
212.9 
268.5 

0.9078 
0.7199 
0.5709 

30 
31 
32 

0.2546 
0.2268 
0.2019 

0.05093 
0.04039 
0.03203 

338.6 
426.9 
538.3 

0.4527 
0.3590 
0.2847 

33 
34 
35 

0.1798 
0.1601 
0.1426 

0.02540 
0.02014 
0.01597 

678.8 
856.0 
1079 

0.2258 
0.1791 
0.1420 

36 
37 
38 

0.1270 
0.1131 
0.1007 

0.01267 
0.01005 
0.007967 

1361 
1716 
2164 

0.1126 
0.08931 
0.07083 

39 
40 
41 

0.08969 
0.07987 
0.07113 

0.006318 
0.005010 
0.003973 

2729 
3441 
4339 

0.05617 
0.04454 
0.03532 

42 
43 
44 

0.06334 
0.05641 
0.05023 

0.003151 
0.002499 
0.001982 

5472 
6900 
8700 

0.02801 
0.02222 
0.01762 

2-14 


PROPERTIES   OF  MATERIALS 


Table  7.    Solid  Copper  Wire 

British  Standard  Wire  Gage;  English  Units 
100  per  cent  conductivity;  density  8.89  at  20  deg  cent 


Gage  No. 

Diameter, 
mils 

Cross-section 

Ohms  per 
1000ft, 
15.6°  C  or 
60°  F* 

Pounds  per 
1000  ft 

Circular  Mils 

Square  Inches 

7-0 

500 

250,000 

0.1964 

0.04077 

756.8 

6-0 

464 

215,300 

0.1691 

0.04734 

651.7 

5-0 

432 

186,600 

0.1466 

0.05461 

564.9 

4-0 

400 

160,000 

0.1257 

0.06370 

484.3 

3-0 

372 

138,400 

0.  1087 

0.07365 

418.9 

2-0 

348 

121,100 

0.09512 

0.08416 

366.6 

0 

324 

105,000 

0.08245 

0.09709 

317.8 

1 

300 

90,000 

0,07069 

0.1132 

272.4 

2 

276 

76,180 

0.05983 

0.1338 

230.6 

3 

252 

63,500 

0.04988 

0.1605 

192.2 

4 

232 

53,820 

0.04227 

0.1894 

162.9 

5 

212 

44,940 

0.03530 

0.2268 

136.0 

6 

192 

36,860 

0.02895 

0.2765 

111.6 

7 

176 

30,980 

0.02433 

0.3290 

93.76 

8 

160 

25,600 

0.02011 

0.3981 

77.49 

9 

144 

20,740 

0.01629 

0.4915 

62.77 

10 

128 

16,380 

0.01287 

0.6221 

49.59 

11 

116 

13,460 

0.01057 

0.7574 

40.73 

12 

104 

10,820 

0.008495 

0.9423 

32.74 

13 

92 

8,464 

0.006648 

1.204 

25.62 

14 

80 

6,400 

0.005027 

1.592 

19.37 

15 

72 

5,184 

0.004072 

1.966 

15.69 

16 

64 

4,096 

0.003217 

2.488 

12.40 

17 

56 

3,136 

0.002463 

3.250 

9.493 

18 

48 

2,304 

0.001810 

4.424 

6.974 

19 

40 

1,600 

0.001257 

6.370 

4.843 

20 

36 

1,296 

0.001018 

7.864 

3.923 

22 

28 

784.0 

0.0006158 

13.00 

2.373 

24 

22 

484.0 

0.0003801 

21.06 

1.465 

26 

18 

324.0 

0.0002545 

31.46 

0.9807 

28 

14.8 

219.0 

0.0001720 

46.54 

0.6630 

30 

12.4 

153.8 

0.0001208 

66.28 

0  4654 

32 

10.8 

116.6 

0.00009161 

87.38 

0.3531 

34 

9.2 

84.64 

0.00006648 

120.4 

0.2562 

36 

7.6 

57.76 

0.00004536 

176.5 

0.  1748 

38 

6.0 

36.00 

0.00002827 

283.1 

OJ090 

40 

4.8 

23.04 

0.00001810 

442.4 

0.06974 

42 

4.0 

16.00 

0.00001257 

637.0 

0.04843 

44 

3.2 

10.24 

0.000008042 

995.3 

0.03100 

46 

2.4 

5.760 

0.000004524 

1,769 

0.01744 

48 

1.6 

2.560 

0.000002011 

3,981 

0,007749 

50 

1.0 

1.000 

0.0000007854 

10,190 

0,003027 

e  i.  *  £et  ^"LF^  £ent  conductivity,  B6o  -  resistance  of  100  per  cent  conductivity  w 
Eahr  (from  table),  Rt  —  resistance  of  wire  of  conductivity  C  at  any  temperature  t  deg 

Rt  -  ~  RQQ  [I  +  0.00223ft  -  60)] 


ire  at  60  deg 
fahr;  then 


WIRE   TABLES 


2-15 


Table  8.    Solid  Copper  Wire 

"  Millimeter  Gage";  Metric  Units  and  Circular  Mile 
100  per  cent  conductivity;  density  8.89  at  20  deg  cent 


Diameter, 
mm 

Cross-section, 
sq  mm 

Ohms  per  Kilo- 
meter, 20°  C 

Kilograms  per 
Kilometer 

Cross-section, 
cir  rn.lls  * 

10.0 
9.0 
8.0 

7.0 
6.0 
5.0 

4.5 
4.0 
3.5 

3.0 
2.5 
2.0      - 

1.8 
1.6 
1.4 

1.2 
1.0 
0.90 

78.54 
63.62 
50.27 

38.48 
28.27 
19.64 

15.90 
12.57 
9.621 

7.069 
4.909 
3.142 

2.545 
2.011 
1.539 

1.131 
0.7854 
0.6362 

0.2195 
0.2710 
0.3430 

0.4480 
0.6098 
0.8781 

1.084 
1.372 
1.792 

2.439 
3.512 
5.488 

6.775 
8.575 
11.20 

15.24 
21.95 
27.10 

698.2 
565.6 
446.9 

342.1 
251.4 
174.6 

141.4 
111.7 

85.53 

62.84 
43.64 
27.93 

22.62 
17.87 
13.69 

10.05 
6.982 
5.656 

155,000 
125,550 
99,200 

75,950 
55,800 
38,750 

31,380 
24,800 
18,990 

13,950 
9,690 
6,200 

5,010 
3,970 
3,040 

2,230 
1,550 

0.80 

0.5027 

34.30 

4.469 

0.70 

0.3848 

44.80 

3.421 

0.60 

0.2827 

60.98 

2.514 

0  50 

0   1964 

87  81 

1   746 

0  45 

0.  1590 

108.4 

1   414 

0.40 

0.1257 

137.2 

1.117 

0  35 

0  09621 

179  2 

0.8553 

0  30 

0  07069 

243.9 

0  6284 

0.25 

0.04909 

351.2 

0.4364 

0.20 

0.03142 

548.8 

0.2793 

0   15 

0.01767 

975.6 

0.  1571 

0   10 

0  007854 

2195 

0.06982 

0.05 

0.001964 

8781 

0.01746 

*  One  square  millimeter  equals  1973.52  circular  mile. 


2-16  PROPERTIES   OF  MATERIALS 

Table  9.    Solid  Copper  Wire;  Ohms  per  Unit  Weight 
A.  W.  G.  or  B.  &  S.  Gage;  English  and  Metric  Units 
100  per  cent  conductivity ;  density  8.89  at  20  deg  cent 


Gage 
No. 

Ohms  per  Pound 

Ohms  per  Kilogram 

0°  C 
32°  F 

20°  C 

68°  F 

50°  C 
122°F 

0°C 

20°  C 

50°  C 

0000 

0.0000705! 

0.00007652 

0.00008554 

0.0001554 

0.0001687 

0.0001886 

000 

0.0001121 

0.0001217 

0.0001360 

0.0002472 

0.0002682 

0.0002999 

00 

0,0001783 

0.0001935 

0.0002163 

0.0003930 

0.0004265 

0.0004768 

0 

0.0002835 

0.0003076 

0.0003439 

0.0006249 

0.0006782 

0.0007582 

1 

0.0004507 

0.0004891 

0.0005468 

0.0009936 

0.001078 

0.001206 

2 

0.0007166 

0.0007778 

0.0008695 

0.001580 

0.001715 

0.001917 

3 

0.001140 

0.001237 

0.001383 

0.002512 

0.002726 

0.003048 

4 

0.001812 

0.001966 

0.002198 

0.003995 

0.004335 

0.004846 

5 

0.002881 

0.003127 

0.003495 

0.006352 

0.006893 

0.007706 

6 

0.004581 

0.004972 

0.005558 

.0.01010 

0.01096 

0.01225 

7 

0.007284 

0.007906 

0.008838 

0.01606 

0.01743 

0.01948 

8 

0.01158 

0.01257 

0.01405 

0.02553 

0.02771 

0.03098 

9 

0.01842 

0.01999 

0.02234 

0.04060 

0.04407 

0.04926 

10 

0.02928 

0.03178 

0.03553 

0.06456 

0.07006 

0.07833 

11 

0.04656 

0.05053 

0.05649 

0.1026 

0.1114 

0.1245 

12 

0.07404 

0.08035 

0.08983 

0.1632 

0.1771 

0.  1980 

13 

0.1177 

0.1278 

0.1428 

0.2595 

0.2817 

0.3149 

14 

0.1872 

0.2032 

0.2271 

0.4127 

0.4479 

0.5007 

15 

0.2976 

0.3230 

0.3611 

0.6562 

0.7121 

0.7961 

16 

0.4733 

0.5136 

0.5742 

1.043 

1.132 

1.266 

17 

0.7525 

0.8167 

0.9130 

1.659 

1.800 

2.013 

18 

1.197 

1.299 

1.452 

2.638 

2.863 

3.201 

19 

1.903 

2.065 

2.308 

4.194 

4.552 

5.089 

20 

3,025 

3.283 

3.670 

6.670 

7.238 

8.092 

21 

4.810 

5.221 

5.836 

10.60 

11.51 

12.87 

22 

7.649 

8.302 

9.280 

16.86 

18.30 

20.46 

23 

12.16 

13.20 

14.76 

26.81 

29.10 

32.53 

24 

19.34 

20.99 

23.46 

42.63 

46.27 

51.73 

25 

30.75 

33.37 

37.31 

67.79 

73.57 

82.25 

26 

48.89 

53.06 

59.32 

107.8 

117.0 

131.8 

27 

77.74 

84.37 

94.32 

171.4 

186.0 

207  9 

28 

123.6 

134.2 

150.0 

272.5 

295.8 

330.6 

29 

196.6 

213.3 

238.5 

433.3 

470.3 

525.7 

30 

312.5 

339.2 

379.2 

689.0 

747.8 

836.0 

31 

497.0 

539.3 

602.9 

1,096 

1,189 

1,329 

32 

790.2 

857.6 

958.7 

1,742 

1,891 

2,114 

33 

1,256 

1,364 

1,524 

2,770 

3,006 

3,361 

34 

1,998 

2,168 

2,424 

4,404 

4,780 

5,344 

35 

3,177 

3,448 

3,854 

7,003 

7,601 

8,497 

36 
37 
38 

5,051 
8,032 
12,770 

5,482 
8,717 
13,860 

6,128 
9,744 
15,490 

11,140 
17,710 
28,150 

12,080 
19,220 
30,560 

13,510 
21,480 
34,160 

39 
40 
41 

20,310 
32,290 
51,340 

22,040 
35,040 
55,720 

24,640 
39,170 
62,290 

44,770 
71,180 
113,200 

48,590 
77,260 
122,800 

54,310 
86,360 
137,300 

42 
43 
44 

81,640 
129,800 
206,400 

88,600 
140,900 
224,000 

99,050 
157,500 
250,400 

180,000 
286,200 
455,000 

195,300 
310,600 
493,900 

218,400 
347,200 
552,100 

WIRE   TABLES 


2-17 


Table  10.    Solid  Aluminum  Wire 

A.  W.  G.  or  B.  &  S.  Gage;  English  Units 
61  per  cent  conductivity;  density  2.70 


Gage 
No. 

Diam- 
eter, 
mils 

Cross-section 

Resistance  at 
20°  C  or  68°  F  * 

Weight  in  Pounds 

Feet  per 
Pound 

Circular 
Mils 

Square 
Inches 

Ohms  per 
1000ft 

Ohms  per 
Mile 

per 
1000  ft 

per 
Mile 

0000 

460.0 

211,600 

0.1662 

0.0804 

0.424 

195 

1027 

5.14 

000 

409.6 

167,800 

0.1318 

0.101 

0.535 

154 

815 

6.48 

00 

364.8 

153,100 

0.1045 

0.128 

0.675 

122 

646 

8.17 

0 

324.9 

105,500 

0.08289 

0.161 

0.851 

97.0 

512 

10.31 

1 

289.3 

83,690 

0.06573 

0.203 

1.073 

76.9 

406 

13.00 

2 

257.6 

66,370 

0.05213 

0.256 

1.353 

61.0 

322 

16.39 

3 

229.4 

52,630 

0.04134 

0.323 

1.706 

48.4 

255 

20.7 

4 

204.3 

41,740 

0.03278 

0.408 

2.15 

38.4 

203 

26.1 

5 

181.9 

33,100 

0.02600 

0.514 

2.71 

30.4 

160.7 

32.9 

6 

162.0 

26,250 

0.02062 

0.648 

3.42 

24.1 

127.4 

41.4 

7 

144.3 

20,820 

0.01635 

0.817 

4.31 

19.1 

101.0 

52.3 

8 

128.5 

16,510 

0.01297 

1.03 

5.44 

15.2 

80.2 

65.9 

10 

101.9 

10,380 

0.008155 

1.64 

8.65 

9.55 

50.4 

104.8 

12 

80.81 

6,530 

0.005129 

2.61 

13.76 

6.00 

31.7 

166.6 

14 

64.08 

4,107 

0.003225 

4.14 

21.9 

3.78 

19.93 

265 

15 

57.07 

3,257 

0.002558 

5.22 

27.6 

2.99 

15.81 

334 

16 

50.82 

2,583 

0.002029 

6.59 

34.8 

2.37 

12.54 

421 

17 

45.26 

2,048 

0.001609 

8.31 

43.8 

1.88 

9.94 

531 

18 

40.30 

1,624 

0.001276 

10.5 

55.3 

1.49 

7.89 

670 

19 

35.89 

1,288 

0.001012 

13.2 

69.7 

1.18 

6.25 

844 

20 

31.96 

1,022 

0.0008023 

16.7 

87.9 

0.939 

4.96 

1,065 

21 

28.46 

810.1 

0.0006363 

21.0 

110.9 

0.745 

3.93 

1,343 

22 

25.35 

642.4 

0.0005046 

26.5 

139.8 

0.591 

3.12 

1,693 

23 

22.57 

509.5 

0.0004002 

33.4 

176.3 

0.468 

2.47 

2,130 

24 

20.10 

404.0 

0.0003173 

42.1 

222 

0.371 

1.961 

2,690 

25 

17.90 

320.4 

0.0002517 

53.1 

280 

0.295 

1.556 

3,390 

26 

15.94 

254.1 

0.0001996 

67.0 

353 

0.234 

1.233 

4,280 

27 

14.20 

201.5 

0.0001583 

84.4 

446 

0.185 

0.978 

5,400 

28 

12.64 

159.8 

0.0001255 

106 

562 

0.147 

0.776 

6,810 

29 

11.26 

126.7 

0.00009953 

134 

709 

0.117 

0.615 

8,580 

30 

10.03 

100.5 

0.00007894 

169 

894 

0.0924 

0.488 

10,820 

31 

8.928 

79.70 

0.00006260 

213 

1127 

0.0733 

0.387 

13,650 

32 

7.950 

63.21 

0.00004964 

269 

1421 

0.0581 

0.307 

17,210 

33 

7.080 

50.13 

0.00003937 

339 

1792 

0.0461 

0.243 

21,700      ' 

34 

6.305 

39.75 

0.00003122 

428 

2260 

0.0365 

0.1929 

27,400 

35 

5.615 

31.52 

0.00002476 

540 

2850 

0.0290 

0.  1530 

34,510 

*  Let  C  =»  per  cent  conductivity, 
cent  (from  table) ,  Rt  =  resistance  ol 

fir 


20  =  resistance  of  61  per  cent  conductivity  wire  at  20  deg 
=  resistance  of  wire  of  conductivity  C  at  any  temperature  t  deg  cent;  then 
61  #20  r 
C 


5  [1  +  0.004«  -  20)1 


2-18 


PROPERTIES   OF  MATERIALS 


Table  11.    Solid  Aluminum  Wire 

A.  W.  G.  or  B.  &  S.  Gage  in  Metric  Units 
61  per  cent  conductivity;  density  2.70;  temperature  20  deg  cent  or  68  deg  fahr  * 


Gage  No. 

Diameter, 

mm 

Cross-section, 

Ohms  per 
Kilometer 

Kilograms  per 
Kilometer 

0000 

11.68 

107.2 

0.264 

289 

000 

10.40 

85.03 

0.333 

230 

00 

9.266 

67.43 

0.419 

182 

0 

8.252 

53.48 

0.529 

144 

1 

7.348 

42.41 

0.667 

114 

2 

6.544 

33.63 

0.841 

90.8 

3 

5.827 

26.67 

1.06 

72.0 

4 

5.189 

21.15 

T.34 

57.1 

5 

4.621 

16.77 

T.69 

45.3 

6 

4.115 

13.30 

2.13 

35.9 

7 

3.665 

10.55 

2.68 

28.5 

8 

3.264 

8.366 

3.38 

22.6 

10 

2.588 

5.261 

5.38 

H.2 

12 

2.053 

3.309 

8.55 

8.93 

14 

1.628 

2.081 

13.6 

5.62 

15 

1.450 

1.650 

17.1 

4.46 

16 

1.291 

1.309 

21.6 

3.53 

17 

1.150 

1.038 

27.3 

2.80 

18 

1.024 

0.8231 

34.4 

2.22 

19 

0.9116 

0.6527 

43.3 

1.76 

20 

0.8118 

0.5176 

54.6 

1.40 

21 

0.7230 

0.4105 

68.9 

T.IT 

22 

0.6438 

0.3255 

86.9 

0.879 

23 

0.5733 

0.2582 

110 

0.697 

24 

0.5106 

0.2047 

738 

0.553 

25 

0.4547 

0.1624 

174 

0.438 

26 

0.4049 

0.1288 

220 

0.348 

27 

0.3606 

0.1021 

277 

0.276 

28 

0.3211 

0.08098 

349 

0  219 

29 

0.2859 

0.06422 

440 

0.173 

30 
31 
32 

0.2546 
0.2268 
0.2019 

0.05093 
0.04039 
0.03203 

555 
700 
883 

0.138 
0.109 
0.0865 

33 
34 
35 

0.1798 
0.1601 
0.1426 

0.02540 
0.02014 
0.01597 

1110 
1400 
1770 

0.0686 
0.0544 
0.0431 

cent 


-  resistance  of  61 
wire  of  conductivity  C 


per  cent  conductivity  wire  at  20  dee 
at  any  tempeiature  t  deg  ;  cent;    the? 


-  20)] 


The  temperature  coefficient  is  approzimate  only. 


WIRE  TABLES 


2-19 


Table  12.    Solid  Steel  Wire 

American  Steel  Wire  Gage;  English  Units 

12.5  per  cent  conductivity;  density  7.78 


Am. 
Steel 
Wire 
Gage 
No. 

Diameter 

Cross-section 

Resistance  at 
20°  C  or  68°  F  * 

Weight  in 
Pounds 

Feet 
per 
Pound 

In. 

Mils 

Circular 
Mils 

Square 
Inches 

Ohms  per 
1000ft 

Ohms  per 
Mile 

per 
1000ft 

per 

Mile 

1/2 

500.0 

250,000 

0.1964 

0.332 

1.752 

662.5 

3499 

.51 

7-0 

490.0 

240,100 

0.1886 

0.346 

1.825 

636.3 

3360 

.53 

15/32 

468.8 

219,800 

0.1726 

0.378 

1.993 

582.4 

3075 

.72 

6-0 

460.0 

211,600 

0.1662 

0.392 

2.07 

560.8 

2961 

.78 

Vl6 

437.5 

191,400 

0.1503 

0.433 

2.29 

507.2 

2678 

.97 

5-0 

430.0 

184,900 

0.1452 

0.449 

2.37 

490.0 

2587 

2.04 

13/32 

406.3 

165,000 

0.1296 

0.503 

2.65 

436.8 

2306 

2.28 

4-0 

393.8 

155,100 

0.1218 

0.535 

2.82 

411.9 

2175 

2.42 

3/8 

375.0 

140,600 

0.1104 

0.590 

3.12 

372.6 

1967 

2.68 

3-0 

362.5 

131,400 

0.1032 

0.631 

3.33 

348.2 

1839 

2.87 

H/32 

343.8 

118,200 

0.09280 

0.702 

3.71 

313.1 

1653 

3.19 

2-0 

331.0 

109,600 

0.08605 

0.757 

4.00 

290.3 

1533 

3.44 

5/16 

312.5 

97,660 

0.07670 

0.850 

4.49 

258.8 

1366 

3.86 

0 

306.5 

93,940 

0.07378 

0.883 

4.66 

249.0 

1315 

4.02 

1 

283.0 

80,090 

0.06290 

1.036 

5.47 

212.2 

1121 

4.71 

9/32 

281.3 

79,100 

0.06213 

.049 

5.54 

209.6 

1107 

4.77 

2 

262.5 

68,910 

0.05412 

.204 

6.36 

182.6 

964.1 

5.48 

V4 

250.0 

62,500 

0.04909 

.328 

7.01 

165.6 

874.5 

6.04 

3 

243.7 

59,490 

0.04665 

.397 

7.38 

157.4 

831.0 

6.35 

4 

225.3 

50,760 

0.03987 

.635 

8.63 

134.5 

710.2 

7.43 

7/32 

218.8 

47,850 

0.03758 

.734 

9.15 

126.8 

669.5 

7.89 

5 

207.0 

42,850 

0.03365 

1.936 

10.22 

113.6 

599.5 

8.81 

6 

192.0 

36,860 

0.02895 

2.25 

11.88 

97.7 

515.8 

10.23 

3/16 

187.5 

35,160 

0.02761 

2.36 

12.46 

93.2 

491.9 

10.73 

7 

177.0 

31,330 

0.02461 

2.65 

13.98 

83.0 

438.4 

12.04 

8 

162.0 

26,240 

0.02061 

3.16 

16.69 

69.6 

367.2 

14.38 

5/32 

156.3 

24,410 

0.01917 

3.40 

17.95 

64.7 

341.6 

15.46 

9 

148.3 

21,990 

0.01727 

3.77 

19.92 

58.3 

307.8 

17.16 

10 

135.0 

18,200 

0.01431 

4.55 

24.0 

48.3 

255.0 

20.70 

V8 

125.0 

15,630 

0.01227 

5.31 

28.0 

41.4 

218.6 

24.15 

11 

120.5 

14,520 

0.01140 

5.71 

30.2 

38.5 

203.2 

25.98 

12 

105.5 

11,130 

0.00874 

7.45 

39.4 

29.5 

155.7 

33.90 

3/32 

93.8 

8,789 

0.00690 

9.44 

49.8 

23.3 

123.0 

42.94 

13 

91.5 

8,372 

0.00658 

9.91 

52.3 

22.1 

117.2 

45.16 

14 

80.0 

6,400 

0.00503 

12.96 

68.5 

17.0 

89.55 

58.97 

15 

72.0 

5,184 

0.00407 

16.01 

84.5 

13.7 

72.53 

72.80 

16 

62.5 

3,906 

0.00307 

21.2 

112.1 

10.4 

54.66 

96.60 

1/16 

62.5 

3,906 

0.00307 

21.2 

112.1 

10.4 

54.66 

96.60 

17 

54.0 

2,916 

0.00229 

28.5 

150.2 

7.73 

40.80 

129.5 

18 

47.5 

2,256 

0.00177 

36.8 

194.2 

5.98 

31.57 

167.2 

19 

41.0 

1,681 

0.00132 

49.4 

261 

4.45 

23.52 

224.4 

20 

34.8 

1,211 

0.00095 

68.5 

362 

3.21 

16.95 

311.5 

21 

31.8 

1,008 

0.00079 

82.3 

435 

2.67 

14.11 

374.4 

1/32 

31.3 

977 

0.00076 

85.0 

449 

2.59 

13.66 

386.5 

22 

28.6 

818 

0.00064 

101.4 

536 

2.17 

11.45 

461.1 

23 

25.8 

666 

0.00052 

124.6 

658 

1.76 

9.31 

567.0 

24 

23.0 

529 

0.00042 

156.8 

828 

1.40 

7.40 

713.5 

25 

20.4 

416 

0.00033 

199.4 

1053 

1.10 

5.82 

907.0 

*  Let  C  ~  per  cent  conductivity, 

#20  =  resistance  of  12.5  per  cent  conductivity  wire  at  20  deg  cent  (from  table), 
Rt  —  resistance  of  wire  of  conductivity  C  at  any  temperature  t  deg  cent;  then 

Rt  „  12-*20  [1  +  0.006(*  -  20)] 


The  temperature  coefficient  is  approximate  only. 


2-20 


PROPERTIES   OF  MATERIALS 


Table  12.  Solid  Steel  Wire— Continued 
American  Steel  Wire  Gage;  English  Units 
12.5  per  cent  conductivity;  density  7.78 


Am. 
Steel 

Diameter 

Cross-section 

Resistance  at 
20°  C  or  68°  F  * 

Weight  in 
Pounds 

Feet 
per 

Wire 
Gage 

No. 

In. 

Mils 

Circular 
Mils 

Square 
Inches 

Ohms  per 
1000  ft 

Ohms  per 
Mile 

per 
1000  ft 

per 

Mile 

Pound 

26 

18.1 

328 

0.00026 

253 

1337 

0.87 

4.58 

1152 

27 

17  3 

299 

0.00024 

277 

1464 

0.79 

4.19 

1261 

28 

16.2 

262 

0.00021 

316 

1669 

0.70 

3.67 

1438 

29 

15.0 

225 

0.00018 

469 

1947 

0.60 

3.15 

1677 

30 

14  0 

196 

0.00015 

424 

2240 

0.52 

2.74 

1925 

31 

13.2 

174 

0.00014 

476 

2510 

0.46 

2.44 

2166 

32 

12.8 

164 

0.00013 

506 

2670 

0.43 

2.30 

2303 

33 

11.8 

139 

0.00011 

596 

3150 

0.37 

1.95 

2710 

34 

10.4 

108 

0.00008 

767 

4050 

0.29 

1.51 

3489 

35 

9.5 

90 

0.00007 

919 

4850 

0.24 

1.26 

4193 

36 

9.0 

81 

0  00006 

1023 

5410 

0.21 

1.13] 

4659 

*  Let  C  «*  per  cent  conductivity, 

£20  =  resistance  of  12.5  per  cent  conductivity  wire  at  20  deg  cent  (from _  table), 
Rt  =»  resistance  of  wire  of  conductivity  C  at  any  temperature  t  deg  cent;  then 

Rt  =  is-yfro  u  +  Oj006(<  _  20)] 

The  temperature  coefficient  is  approximate  only. 

COPPER-CLAD  STEEL  WIRE.  This  wire  consists  of  a  steel  core  and  a  concentric 
coat  of  copper  permanently  welded  thereto.  It  is  used  chiefly  for  long-span  transmission 
and  telephone  wire.  It  is  made  in  several  grades,  which  differ  in  the  relative  amounts  of 
steel  and  copper.  The  grades  are  designated  by  the  corresponding  conductivity  expressed 
as  percentages  of  the  Annealed  Copper  Standard:  e.  g.,  40  per  cent  grade  has  a  conductivity 
of  40  per  cent. 

Table  13.    Copper-clad  Steel  Wire 

A.  W,  G.  or  B.  &  S.  Gage;  English  Units 

40  per  cent  conductivity;  density  8.26 


Gage 
No. 

Diam- 
eter, 
mils 

Cross-section 

Resistance  at 
23.9°  C  or  75°  F  * 

Weight  in 
Pounds 

Feet 
per 
Pound 

Circular 
Mils 

Square 
Inches 

Ohms  per 
1000ft 

Ohms  per 
Mile 

per 
1000ft 

per 

Mile 

0000 

460.0 

211,600 

0.  1662 

0.123 

0.649 

595 

3140 

1.68 

000 

409.6 

167,800 

0.1318 

0.154 

0.813 

471 

2490 

2.12 

00 

364.8 

133,100 

0.1045 

0.195 

1.03 

374 

1970 

2.67 

0 

324.9 

105,500 

0.08289 

0.246 

1.30 

297 

1570 

3.37 

1 

289.3 

83,690 

0.06573 

0.310 

1.64 

235 

1240 

4.26 

2 

257.6 

66,370 

0.05213 

0.390 

2.06 

186 

982 

5.38 

3 

229.4 

52,630 

0.04134 

0.492 

2.60 

148 

781 

6.76 

4 

204.3 

41,740 

0.03278 

0.622 

3.28 

117 

618 

8.55 

5 

181.9 

33,100 

0.02600 

0.782 

4.13 

92.9 

491 

10.76 

6 

162.0 

26,250 

0.02062 

0.987 

5.21 

73.7 

389 

13.57 

7 

144.3 

20,820 

0.01635 

1.25 

6.60 

58.5 

309 

17.09 

8 

128.5 

16,510 

0.01297 

1.57 

8.29 

46.4 

245 

21.6 

9 

114.4 

13,090 

0.01028 

1.98 

10.5 

36.8 

194 

27.2 

10 

101.9 

10,380 

0.008155 

2.50 

13.2 

29.2 

154 

34.2 

11 

90.74 

8,234 

0.006467 

3.15 

16.6 

23.1 

122 

43.3 

12 

80.81 

6,530 

0.005129 

3.97 

21.0 

18.3 

96.6 

54.6 

13 

71.96 

5,178 

0.004067 

5.00 

26.4 

14.6 

77.1 

68.5 

14 

64.08 

4,107 

0.003225 

6.31 

33.3 

11.5 

60.7 

87.0 

*  Let  (7  »  per  cent  conductivity, 

-Kw. 9  *=  resistance  of  40  per  cent  conductivity  wire  at  23.9  deg  cent  (from  table), 
Rt  =  resistance  of  mre  of  conductivity  C  at  temperature  t  deg  cent;  then. 

Rt  =  40  ^23-9  [1  +  0.00432(4  -  23.9)] 
c 

The  temperature  coefficient  ia  approximate  only. 


DIELECTRIC  PROPERTIES 


2-21 


ALLOY  WIRES  OF  HIGH  TENSILE  STRENGTH.  Copper  alloys  having  a  low  con- 
ductivity but  a  tensile  strength  from  50  to  100  per  cent  greater  than  that  of  copper  are 
sometimes  used  where  strength  or  hardness  is  a  primary  requisite,  as  in  long  spans  of 
small  wires  or  for  trolley  wires. 

TENSILE  BREAKING  LOAD.  The  tensile  strength  in  pounds  for  solid  wires  from 
1/16  to  1/2  in.  in  diameter  is  given  in  Table  14. 

Table  14.    Breaking  Load  for  Solid  Wires  in  Pounds  per  Wire 


Gage  No. 
A.W.G. 
or  B.  &  S. 

Diameter 

Hard-drawn 
Copper 
(A.S.T.M.)  * 

Hard-drawn 
Aluminum 
(23,000  to 
33,300  ib 
per  sq  in.) 

Copper-clad 
Steel, 
40  per  cent 
Grade 

Steel 
(100,0001b 
per  sq  in.)  f 

In. 

Mils 

V2 

500 

9310 

4520 

11,400 

19,640 

0000 

460 

8140 

3820 

10,000 

16,620 

Vl6 

437 

7500 

3460 

9,250 

15,030 

000 

410 

6720 

3030 

8,300 

13,180 

3/8 

375 

5800 

2540 

7,150 

11,040 

00 

365 

5540 

2400 

6,850 

10,450 

0 

325 

4520 

1910 

5,700 

8,289 

5/16 

312 

4220 

1770 

5,400 

7,670 

1 

289 

3680 

1530 

4,800 

6,573 

2 

258 

3000 

1240 

4,000 

5,213 

V4 

250 

2830 

1170 

3,780 

4,909 

3 

229 

2420 

1000 

3,200 

4,134 

4 

204 

1950 

810 

2,600 

3,278 

3/16 

187 

1680 

693 

2,300 

2,761 

5 

182 

1570 

655 

2,200 

2,600 

6 

162 

1270 

532 

1,800 

2,062 

7 

144 

1020 

432 

1,450 

1,635 

8 

129 

822 

351 

1,200 

1,297 

Vs 

125 

780 

335 

1,150 

1,227 

9 

114 

660 

287 

975 

1,028 

10 

102 

528 

234 

800 

816 

11 

91 

423 

191 

650 

647 

12 

81 

337 

155 

510 

513 

13 

72 

268 

126 

410 

407 

14 

64 

213 

103 

330 

323 

Vl6 

62 

203 

98 

310 

307 

*  Tensile  strength  in  pounds  per  square  inch  ranging  from  49,000  for  No.  0000  to  66,200  for 
No.  14;  see  below. 

t  For  wires  having  a  tensile  strength  of  S  pounds  per  square  inch,  multiply  by  5/100,000.  The 
tensile  strength  of  steel  varies  from  60,000  to  225,000  Ib  per  sq  in. 


INSULATING  MATERIALS 

By  W.  R.  Dohan 

4.  DIELECTRIC  PROPERTIES 

Dielectric  Constant.  The  dielectric  constant,  K,  of  an  insulating  material  is  the  ratio  of 
the  capacitance  of  an  electrode  system  using  the  material  as  a  dielectric  to  its  capacitance 
with  a  vacuum  dielectric.  When  the  material  has  appreciable  losses  so  that  the  parallel 
capacitance,  Cp  (determined  by  comparison  with  a  standard  capacitor  and  a  parallel  con- 
ductance, <2),  differs  significantly  from  the  effective  series  capacitance  Ca  (determined  by 
comparison  with  a  standard  capacitor  and  a  series  resistance,  R),  it  is  standard  to  use 
the  value  of  Cp  in  computing  the  dielectric  constant.  These  capacitances  are  related  by 
the  following  formula: 


where  8  =  the  loss  angle  or  phase  difference. 
D  —  the  dissipation  factor. 


2-22  PROPERTIES  OF  MATERIALS 

Since  the  dielectric  constant  may  vary  considerably  with  frequency  or  temperature  the 
conditions  of  measurement  should  always  be  stated. 

Phase  Difference  or  Loss  Angle.  The  difference  between  the  theoretical  90  electrical 
degrees  phase  advance  of  the  current  through  a  perfect  capacitor  and  the  actual  angle  of 
phase  advance,  6,  of  the  current  through  the  dielectric  material  is  known  as  the  phase 
difference,  or  loss  angle,  5. 

Dissipation  Factor.  For  convenience  in  reference  and  calculation  the  tangent  oi^the 
loss  angle,  5,  has  been  assigned  the  standard  designation  "dissipation  factor"  and  given 
the  symbol  D. 

Power  Factor.  The  power  factor  of  a  dielectric  material  is  the  ratio  of  the  power  loss 
in  the  material  to  the  product  of  the  applied  voltage  and  current.  The  power  factor  there- 
fore is  equal  to  the  cosine  of  the  angle  of  phase  advance,  0,  and  to  the  sine  of  the  comple- 
mentary angle,  5.  The  sine  and  tangent  of  a  small  loss  angle  are  very  nearly  the  same,  so 
that  the  power  factor  is  substantially  equal  to  the  dissipation  factor  for  values  less  than  0.1. 

Loss  Factor.  The  product  of  the  dielectric  constant  and  the  dissipation  factor  (or 
power  factor  if  the  value  is  less  than  0.1)  is  known  as  the  loss  factor. 

Power  Loss.  The  power  loss  in  a  dielectric  may  be  considered  to  take  place  in  a  ficti- 
tious shunt  or  series  resistance,  depending  on  the  material,  the  applied  frequency,  and 
the  temperature.  For  a  series  resistance,  R,  the  dissipation  factor  is 

D  =  tan  5  =  R2irfC8  =  Power  factor 
For  a  parallel  conductance,  G,  or  shunt  resistance,  r,  the  dissipation  factor  is 

D  =  tan  5  =  rt    ,.,    =    .    ,„    ===  Power  factor 
2-irfCp       rZ-rrfCp 

In  the  above  formulas  Ct  and  Cp  are  substantially  equal  for  dissipation  factors  less  than  0.1. 
The  power  loss,  P,  in  a  parallel  resistance  is 

m 

watts 

The  power  loss  in  a  small  series  resistance  is 

P  =  &•  [27r/(7fl]2  R  =  ZPSTrfCsD  watts 

Writing  Cs  =  Cp  —  C  in  terms  of  K,  area  in  square  centimeters  A,  thickness  in  centi- 
meters i,  and  the  dielectric  coefficient  for  free  space  8.854  X  10  ~12, 

C  =  8.854  X  lO-14-^  •  ~ 
and  AK 

P  =  0.555  X  10-12  •  &•  —  -f-D  watts 
t 

which  may  be  rewritten  in  terms  of  volume  in  cubic  centimeters,  V, 

(E\z 
-  }    •  V-f-  (K-D}  watts  per  cm3  per  cycle  per  (volt  per  cm)2 


Thus  the  power  loss  is  seen  to  (1)  increase  as  the  square  of  the  voltage  gradient 
(2)  increase  with  the  volume  of  material  in  the  field;  (3)  increase  with  frequency  if  (K.-D] 
is  constant;  (4)  increase  with  the  product  (K-D},  the  loss  factor.  Shunt  resistance  or 
conductance  is  often  the  most  important  cause  of  losses  at  frequencies  below  10  kc,  since 
the  impedance  of  the  capacitor  is  relatively  high  compared  to  the  shunt  resistance.  At  a 
frequency  of  100  kc  the  impedance  will  be  much  lower  compared  to  the  shunt  resistance, 
and  at  1000  ke  the  shunt  resistance  of  good  dielectric  materials  is  extremely  high  com- 
pared to  the  impedance.  In  the  frequency  range  from  1000  kc  to  1000  Me  the  dielectric 
constant  and  dissipation  factor  of  good  non-polar  dielectric  materials  vary  but  slightly, 
thus  indicating  a  constant  loss  per  cycle. 

Polar  and  Non-polar  Materials.  A  polar  substance  is  characterized  by  a  permanent 
unbalance  in  the  electric  charges  within  a  molecule.  This  unbalanced  charge  system  is 
known  as  a  "dipole"  and  tends  to  turn  in  an  electric  field.  In  liquids  and  soft  solids  which 
are  polar,  there  is  a  free  rotation  of  the  dipoles  at  certain  temperatures  and  applied  fre- 
quencies, causing  a  very  high  loss. 

Non-polar  materials  have  no  permanent  charge  unbalance:  though  the  molecule  may  be 
distorted  by  an  applied  electric  field,  no  tendency  to  rotate  exists.  Non-polar  substances 
therefore  are  free  of  sharp  loss  peaks  as  the  temperature  or  frequency  is  varied,  any  changes 
in  dielectric  constant  and  power  factor  occurring  gradually.  Whether  a  substance  is  polar 
or  non-polar  can  usually  be  predicted  from  its  chemical  structure.  Most  hydrocarbons 


DIELECTRIC  PROPERTIES  2-23 

are  non-polar  and  hence  are  numbered  among  the  best  dielectrics,  e.g.,  polyethylene, 
polystyrene,  mineral  wax,  and  oil. 

Dielectric  Absorption  in  Solids.  A  pure  capacitance  may  be  completely  charged  or 
discharged  almost  instantaneously  if  the  resistance  in  the  external  circuit  is  small.  When 
the  capacitor  contains  an  imperfect  dielectric,  current  continues  to  flow  and  the  charge 
increases  for  a  considerable  period.  The  current  slowly  approaches  a  final  value  fixed  by 
the  insulation  resistance  of  the  dielectric.  For  this  reason,  insulation  resistance  readings 
are  taken  after  the  voltage  has  been  applied  for  a  standard  time  of  1  minute. 

When  the  capacitance  is  discharged  through  a  small  resistance,  a  portion  of  the  total 
charge  will  be  instantaneously  dissipated;  this  has  been  called  the  "free  charge."  If  the 
circuit  is  opened,  the  capacitor  will  be  found  to  have  another  and  smaller  charge  known  as 
the  "residual"  or  "bound"  charge.  Sometimes  this  process  may  be  repeated  two  or  three 
times. 

When  an  absorbing  dielectric  is  subjected  to  an  alternating  electric  field,  the  maximum 
charge  is  greater  than  the  free  charge  but  less  than  the  total  charge  in  a  d-c  field.  The 
measured  value  of  the  dielectric  constant  decreases  with  increasing  frequency,  approach- 
ing a  value  corresponding  to  the  "free  charge."  The  instantaneous  charge  is  not  in  phase 
with  the  applied  a-c  voltage,  and  a  loss  of  energy  results,  heating  the  dielectric.  The 
maximum  loss  occurs  at  a  frequency  which  is  equal  to  the  reciprocal  of  the  time  constant 
of  the  charge  or  discharge  current-time  curve.  When  the  frequency  is  increased  until  the 
time  constant  for  one  cycle  is  very  short  compared  to  the  charging  time  constant  with 
direct  current,  practically  no  loss  remains. 

The  exact  mechanism  of  absorption  loss  is  not  known,  although  many  theories  have 
been  suggested,  such  as  surface  charges  in  non-homogeneous  dielectrics  (Maxwell  and 
Wagner) ,  space  charges  (Whitehead  and  Joffe) ,  and  dipoles  (Debye) . 

Insulation  Resistance.  Insulation  resistance  is  the  ratio  of  the  applied  d-c  voltage  to 
the  resultant  current  flowing,  after  1  minute  of  voltage  application,  between  two  elec- 
trodes embedded  in,  or  making  contact  with,  the  dielectric.  The  nature  of  the  specimen 
determines  whether  the  value  represents  principally  surface  resistance  or  volume  resist- 
ance: when  thin  specimens  with  studs  or  bolts  as  electrodes  are  measured  at  high  humid- 
ities, the  value  is  more  nearly  representative  of  surface  resistance;  when  electrodes  of  large 
area  are  applied  to  the  faces  of  a  slab  and  measured  at  low  humidities  the  result  is  more 
nearly  representative  of  volume  resistance. 

Surface  Resistivity.  Surface  resistivity  is  the  resistance  between  opposite  edges  of  a 
square.  Since  the  resistance  of  the  body  of  a  material  is  always  in  parallel  with  the  surface 
resistance,  the  latter  is  measurable  only  when  the  volume  resistance  is  much  greater  than 
the  surface  resistance,  e.g.,  under  conditions  of  high  humidity  and  large  ratio  of  surface 
to  volume  in  the  electric  field. 

The  degree  of  surface  contamination  is  an  important  factor  since  perspiration  or  other 
surface  contaminant  dissolves  in  the  condensed  moisture  layer  and  increases  its  conduc- 
tivity. While  a  layer  of  pure  water  0.1  micron  thick  would  result  in  a  surface  resistivity 
of  about  3  X  1010  ohms  at  room  temperature  (a  value  commensurate  with  most  wetted 
insulating  materials  at  humidities  above  90  per  cent),  a  mere  trace  of  salt  on  the  surface 
would  reduce  the  resistance  by  a  factor  of  10  ~3. 

The  temperature  coefficient  of  surface  resistivity  is  negative. 

Volume  Resistivity.  Volume  resistivity  is  the  resistance  between  opposite  faces  of  a 
centimeter  cube  after  the  surface  leakage  is  eliminated;  it  is  expressed  in  ohm-centimeters. 
Volume  resistivity  is  calculated  from  the  resistance  between  two  electrodes,  one  of  which 
is  completely  surrounded  by  a  guard  electrode  maintained  at  the  same  potential.  The 
current  due  to  the  surface  resistance  flows  through  the  guard  circuit  and  does  not  influence 
the  value  of  current  in  the  guarded  electrode  circuit. 

Volume  resistivity  has  a  negative  temperature  coefficient  and  often  is  found  to  have  a 
negative  voltage  coefficient.  Some  materials,  especially  those  of  a  fibrous  nature,  exhibit 
changes  in  resistance  with  polarity  and  the  resistivity  may  change  with  time,  owing  to 
polarization  or  to  water  migration.  This  is  known  as  the  "Evershed  effect." 

Dielectric  Strength,  of  Solids.  The  dielectric  strength  of  a  material  is  the  maximum 
potential  that  unit  thickness  can  withstand  without  breakdown.  The  value  obtained 
will  depend  upon  sample  thickness,  temperature,  applied  frequency,  wave  form,  electrode 
form,  area  and  heat  conductivity,  the  surrounding  medium,  and  the  rate  and  total  time 
of  voltage  application. 

In  order  that  test  values  may  be  comparable,  the  American  Society  for  Testing  Mate- 
rials has  standardized  these  variables  for  specific  classes  of  materials.  The  dielectric 
strength  values  for  a  given  class  of  material,  e.g.,  molded  thermosetting  plastics,  are 
therefore  comparable  but  do  not  bear  a  direct  relation  to  the  values  for  a  different  class 
of  material,  e.g.,  mica  or  oil. 


2-24  PROPERTIES  OF  MATERIALS 

The  general  effect  of  increase  in  thickness  is  to  raise  the  total  breakdown  voltage :  the 
increase  for  solid  dielectrics  is  nearly  linear  for  small  thicknesses  at  room  temperature  but 
very  much  less  than  linear  for  large  thicknesses  or  higher  temperatures.  Elevation  of 
temperature  invariably  reduces  the  dielectric  strength. 

The  peak  value  of  the  60-cycle  a-c  breakdown  voltage  usually  is  less  than  that  of  the 
d-c  breakdown  voltage.  The  breakdown  voltage  decreases  with  increasing  frequency, 
the  rate  depending  on  the  loss  factor  of  the  material.  Dielectric  strength  at  1  megacycle 
may  be  as  low  as  25  per  cent  of  the  60-cycle  value. 

When  an  electrode  is  sharply  curved,  the  potential  gradient  at  the  surface  is  raised 
according  to  the  laws  of  electrostatics,  and  a  reduction  in  breakdown  voltage  results.  If 
the  electrode  area  is  increased,  the  probability  of  a  weak  dielectric  spot  within  the  electrode 
area  is  likewise  increased,  and  the  average  dielectric  strength  is  reduced. 

The  effect  of  time  on  breakdown  voltage  is  best  stated  by  Peek's  equation: 

g  =  £0(1  -  arty 

where   g  —  dielectric  strength  at  time  t  seconds. 
gQ  —  dielectric  strength  at  infinite  time. 
a  —  a  constant. 

Both  go  and  a  vary  with  temperature  and  thickness.    The  formula  is  unsatisfactory  for 
times  less  than  0.01  sec  or  for  very  long  periods.  * 

The  general  effect  of  placing  dielectrics  in  series  is  to  decrease  the  a-c  breakdown  voltage, 
since  the  voltage  divides  in  inverse  proportion  to  the  dielectric  constants  if  the  resistivities 
are  high.  Laminated  structures,  unless  bonded  with  a  medium  of  the  same  dielectric 
constant  or  impregnated  throughout  with  some  medium,  tend  to  have  a  lower  dielectric 
strength  than  that  of  an  equivalent  thickness  of  homogeneous  material.  The  presence  of 
air  or  gas  between  laminae  causes  a  pronounced  reduction  in  dielectric  strength. 

Flashover.  Flashover  is  an  insulation  failure  by  discharge  between  the  electrodes  over 
the  surface  of  an  insulator.  Sometimes  the  insulator  is  permanently  damaged  by  the 
nashover.  In  a  uniform  electric  field  the  nashover  voltage  at  low  relative  humidities 
approaches  the  dielectric  strength  of  air  as  a  limit.  Increase  in  humidity  causes  a  surface 
moisture  film  to  form  on  the  insulator,  reducing  the  flashover  voltage.  Substances  which 
are  wetted  by  water  form  a  more  or  less  continuous  film,  and  the  nashover  values  are 
somewhat  erratic,  falling  rapidly  up  to  about  50  per  cent  relative  humidity  and  then  more 
slowly.  Non-wetted  substances  such  as  waxes,  on  which  the  moisture  condenses  in  drop- 
lets, show  an  almost  linear,  and  quite  consistent,  decrease  in  nashover  with  increase  in 
humidity.  Higher  temperatures  and  lower  pressures  both  reduce  the  density  of  the  air: 
therefore  both  factors  decrease  the  nashover  value.  The  flashover  voltage  at  high  humid- 
ities may  be  from  three-quarters  to  one-half  of  the  sparkover  voltage  in  the  absence  of 
the  insulator. 

Arc  Resistance.  The  power  arc  following  a  flashover  or  the  breaking  of  contacts  over 
the  insulator  surface  subjects  the  surface  to  extreme  heat,  to  chemical  action,  and  to 
deposition,  of  electrode  material.  Where  exposure  of  the  insulator  to  arcs  cannot  be 
avoided,  it  is  important  to  know  the  degree  of  resistance  to  be  expected.  Glass,  mica,  and 
ceramic  materials  are  quite  resistant  and  become  permanently  conductive  only  by  deposit 
of  electrode  material.  Organic  materials  fail  by  carbonizing  and  become  permanently 
conductive  even  with  intermittent  arcing.  Unfortunately,  the  phenolic  materials  are  very 
poor  in  this  respect,  failing  in  a  few  seconds  when  tested  according  to  ASTM  method 
D495.  Plastics,  such  as  polystyrene,  which  liberate  volatile  monomers,  tend  to  blow  the 
arc  from  the  surface  and  fail  in  about  60  sec.  Shellac,  hard  rubber,  and  vulcanized  fiber 
are  moderately  arc  resistant.  Vinyl  plastics  and  methacrylates  likewise  are  somewhat 
resistant.  The  melamine  resins  are  outstanding  as  a  class:  glass  cloth  melamine  laminates 
last  over  180  sec.  Cellulose  acetates  and  ethyl  cellulose  may  be  formulated  to  exceed 
180  sec  resistance.  Non-refractory  cold-molded  compounds  and  glass-bound  mica  may 
last  from  180  to  400  sec.  Refractory  cold-molded  compounds  last  even  longer,  approach- 
ing the  ceramics.  Arc-resistant  varnishes  are  available  which  considerably  improve  the 
rating  of  phenolic  materials. 

Test  Methods.  Methods  of  testing  electrical  insulating  materials  have  been  standard- 
ized by  the  American  Society  for  Testing  Materials  (1916  Race  St.,  Philadelphia  3,  Pa.). 
A  few  of  the  more  important  standards  are  listed  below : 

Dielectric  Constant  and  Power  Factor ASTM  D150 

Dielectric  Strength ASTM  D149 

Insulation  Resistance ASTM  D257 

Arc  Resistance [""   ASTM  D495 

Reference  should  be  made  to  the  current  Index  to  ASTM  Standards  for  further  information. 


SOLID   DIELECTRIC  MATERIALS  2-25 


5.  SOLED  DIELECTRIC  MATERIALS 

Table  1  lists  the  important  physical  and  electrical  properties  of  solid  materials  useful 
for  electrical  insulation ;  it  is  followed  by  a  list  of  trade  names  and  additional  information. 
The  properties  given  are  not  intended  to  be  design  values  but  to  illustrate  the  range  of 
properties  to  be  expected  in  a  given  class  of  material  or  a  value  typical  of  the  class. 


2-26 


PROPERTIES  OF  MATERIALS 


Physical  Properties  at  25°  C 


Table  1.    Properties  of  Solid 


Density, 
cm3 

Tens. 

Sir., 
Ib 
in.2 

xio3 

Comp. 
Str., 
Ib 
in.2 

xio3 

Mod. 
of 
Elas- 
ticity 

xio6 

Flex. 
Str., 
Ib 
in.2 

xio3 

Coef. 
Lin. 
Ther. 
Exp. 
per  °C 

xio~6 

Ther. 
Cond. 
per°C 

xio~4 

(Note 
A) 

Max. 
Oper. 
Temp., 
°C 
(Note 
B) 

%H20 
Absorp- 
tion in 
24  Hr 

Material 
(See  text  also) 
(Note  C) 

1.3-1.5 
2.6 
2.7 
2.7 
1.05 
1,21 
1.5 
0.5-0.8 
1.01-1.09 
1-1.17 
0.9-1.07 
1.2-1.3 
0.96 
1.32-1.39 
1.35-1.6 

5-6 

8.5 
10 
7.5 
Melt,  pt 
10.0 
9-11 
0.3-0.4 
Melt.  pt. 
Melt.  pt. 
Melt.  pt. 
5-11 
Melt.  pt. 
4.5-10 
4-10 

23.0 
75 
85 
65 
250-325° 
20-22 
38-44 

025-0.4 
15 

5 

9.0 
20 
22 
18 
3 
15-20 
12-16 

90-100 
6.9 
73 
7.5   . 

44 
54 
30 

5.0 
60 

60 
60 

60-80 
1000 
1000 
1000 
180? 
107 
148 
260 

0.3-0.4 
0.00-0.08 
0.00-0.05 
0.00-0.08 

Low 
0.08 
0.2-0.6 

Very  high 

Allyl  resin,  cast        .   . 

Alsimao-  *  No.  35  

«         No.  196  

No.  211  

C 
0.65 
1.2 

Amber        

Aniline  formaldehyde  resin  
Same  glass  mat    .   . 

'"V.45 
14-20 

Asbestos  paper  (dry)  
Asphalt  (native)  

57-87°  C 
27-122°  C 
27-162°  C 
26-33 
60.5-43.5 
5.1-27 
25 

3;  flash  pt 
3;  flash  pt 
0.7-1 
°C 
0.5-0.57 
02-0.39 

200-320° 
.  175-290° 
12-17 

C 
C 
50-110 

"      petroleum  

"      blown  

3-4 

125 

0.05-0.07 

Bakelite  *  resin,  pure  

Beeswax,  yellow  

10-18 
9 

80 
120-160 

7-14 
2.3-3.2 

Casein  plastic  

3.1-5.1 

45 

Celluloid  *  clear 

Cellulose,  dry  

1.28-1.32 
1.25-1.37 
1.25-1.56 
1.25-1.4 

1.14-1.22 
1.17-1.22 
0.91-0.92 
1.07 
ZO-2.4 

7-12 
4-8 
2.5-9 
3-11 

2.7-8.0 
2.8-6.0 

Melt.pt. 

"       acetate,  film  

10-30 
11 
11-27 

7.5-2ZO 

0.15-03 
0.26 
0.1-0.4 

0.1-035 

3.5-10 

120-160 
100 
80-160 

110-160 
120-190 

5.4-6.5 
5.3-8.7 
5.0-9.0 

4.5-7.8 
4-8 

50-70 
50-70 
60-70 

60-75 
60-75 

2.0-7.5 
2.0-4.0 
2.0-7? 

1.6-2.6 
1.0-1.7 

Nil 
0.3 
0.05-5.0 

"           "      transparent  sheet 
"           "      pigmented  sheet 
"           "      molded,  gen.  pur- 
pose 
Cellulose  acetate,  butyrate,  molded 
"        propionate  

5-13 

2.0-13 
4.8-10 

65-75°  C 

Ceresin  wax  

13 
8-10 

100 
1200 

Cerex*  

2-4 

40-50 

7 

1.1-2.5 

30 

Cordierite  ceramic 

Electrose,*  black  

2.5-3.8 
1.05-1.2 
1.01-1.18 
1.1-1.3 
>\3 
>1.05 
2.3-2.9 
2.9-5.9 

10 

20-30 

300 
55 
50-90 
35-60 
125 
125 

Very  low 
1.5-3.0 
1.0-2.0 
1.3-2.0 
<60 
<65 
Nil 
Nil 
Nil 
Nil 
Nil 
<0.01 
Nil 
0.03 
0.003-0.1 
0.06-0.1 

Enamel,  vitreous  

0.4-3.0 
2.7-8.0 
0.5-3.0 
>6.0 
>6.0 
2-5 
3-6 

>0.2 

Ethyl  cellulose,  non-rigid  

8-20 

0.1-0.4 
0.1-0.2 

4-12 

'>J3'" 
>12 

100-140 
120-180 
27 
27 
7.5-11 
8-10.5 

5.6 
3.8-6.2 
4-6 
4-6 
17-25 
14-20 

"          "       rigid,  gen.  pur  
"          "        cast 

>30 
>20 
10-30 
6-10 

Fiber,  bone,  vulcanized 

"      commercial.  .  ,  

8.5-13 
8.7-13 

Glass,  crown  (lime)  

"     flint  (lead) 

8.9 
3.2 
33 
3.2 
0.78 
8 
5-10 
6.4 

17 
27 
24 

"     commercial  plate 

2.25 
Z23 
2.10 
2.18 
3.44 
2.75-3.5 
3.8-3.9 

40 

8.86 

"      Pyrex,*  chem.  resistant  
"         "      elec  #774 

10 
7 

500 
450 
800 
300 
300-400 
300-400 

200 

"     Multiform  *  #  707 

"     Vycor  *  790 

5-8 
6-10 
>6 

0.57 
Melt,  pt 
1.99 
6.0 

22-40 
30-45 
>20 

8 
8 
8 

15-20 
14-19 
13-19 

13 
6-12 
83 

Glass-bonded  mica,  gen.  purpose  .  . 
"         "        "     low  loss  grade  . 
"        "        "     injection 
molded 
Gummon  * 

0.96 

190°  C 
1.56 
100 

200 

4.8 

Gutta  percha 

Hemit  * 

2.5-2.8 
1.83-1.92 
2.5-2.7 
2.8 
Z3 
2.66 
13-1.4 
1.01-1.09 

15.0 

7.0 

173 

1000 

Nil-0.1 

Isolantite  * 

20-30 
20 
20 
96 
24-29 

19.2 
50 
30 

1000 
1200 
1100 

Lava* 

2.0 
2.5 
7.2 
7-11.5 
Nil 

8 
9 
103 
15-19 

83 
2.9 
8.1 
21-24 

1.5 
2.5 
0.01 
0.05-4.2 
ca.60 

"     (Grade  I) 

«     (Grade  A) 

Lavite  * 

0.7-1.4 

ca.  15 

70 
ca.  1000 

Lignin,  sheet  

Magnesium  oxide,  comp.  powd.,  dry 
Marble  blue 

2.6-2.84 
135-1.4 
1.45-1.55 
1.4-1.5 
1.9-1.95 
1J-Z2 

'LS' 

6-7.5 
3.6-7 
16-29 
5-4 

83-213 

12-13 

10-16 

71-91 

0.26 
0.4-1.9 
0.6-1.8 
<1.5 
1.5-3.0 
0.1 

"       white 

Masonite  die  stock  *  
Melamine,  alphacellulose  filled  
"        chopped  rag  filled  

22-25 
23 
38-60 
20 

9-14 
10.5-14 
31-62 
7.5-93 

98 
98 

1.6 

20-45 

125 

"        mineral  filled  

"Trademark  names. 

SOLID  DIELECTRIC  MATERIALS 


2-27 


Dielectric  Materials 


Electrical  Properties  at  25°  C 


Dielectric  Constant 
(Note  D) 

Power  Factor  % 
(Not*  D) 

Volume  Resistivity 
(Note  E) 

Surface  Resistivity, 
ohms  at  20-25°  C 

Dielectric  Strength 
(Note  F) 

Refer- 
ence, 
See 
page  2-32 

Freq. 
less  than 
2kc 

300  to 
2000  kc 

Freq. 
less  than 
2kc 

300  to 
2000  kc 

Ohm-cm 
at 
20-25°  C 

Temp. 
Coef. 
20- 
30°  C 

Relative  Humidity 

Thick- 
ness, 
mils 

Volts 
per  mil 

30% 

50% 

90% 

3.75-4.0 
6.5 
6.3 

3.5 
6.2 
6.0 
5.8 

1.5 
0.3 
0.14 

5.6 
0.2 
0.08 
0.02 
0.513 
0.66 
1.0-1.3 

IG^-IO14 
>1014 
>1014 
>IOU 
5X1016 
1012 
1013 

125 
250 
250 
250 

450 
225 
240 
240 
1400 
600 
450-600 
100 
25-50 

28,  33,  39 
39 
39 
39 
1,16 
39 
32,39 
2,7 
8,  12,  17 
8,  12,  17 
8,  12,  17 
1,12,26 
1,5,8 
26,33 
1,12,33 
12 
39 
12,  28,  39 
39 
40 

32,40 
39 
1,5,8 
33 
19,  21,  39 
1,12 
16 
40 
32,40 
31 
1,5,12,41 
1,5,12,41 
15,  23,  27 
15,23,27 
1,16 
15  " 
15 
27,39 
27,39 
39 
39 
39 

J2 
,  12 
,  12 

39 
,  16 
,  16 
9 
39 
39 
39 
0 
,  5,  7,  12 
,5,7,  12 
39 
40 
40 
32,40 
40 

2.86 
3.73 
4.6 
2.7 
2.7 
3.1 
3.1 

6X1016 

3X1012 
10n-1012 
3.1X109 

3.6 
4.5-5.2 

0.23 
2.0 

125 
62 
47 
90 

2.29 
2.29 

6.1X1014 
6.1X1014 
2X1016 
2X1015 
1.1X1010 

4.5 

0.2 
1.63 

5,19 

2.6 
16.0 

4X1016 
7X1014 

8X1015 
6X1014 
5X109 
5X1010 

8XI014 
5X1014 

250-700 
250 
160-700 
300-700 

2.88 

3.2 
6.15 
6.8 

2.94 

125 
125 

6.7-7.3 
3.9-7.5 
4.3-4.8 
4.5-6 
12 
4.5-6.2 

3.6-6.4 
3.6-3.8 

2.2 
2.7 

6.2-14.4 

7.4-9.7 

2X1010 
1X109 
I014 
5X1010 

1.8 

8X1010 

2X109 

3.3-4.0 
3.3-5 

4.9 
4.0-5.2 

3.0-6.2 
3.3-3.5 

2.5 
2.7 
5-6 

2.0-3.0 
3-8 

15.3 
1.5-4 

1-6 
0.4-1.4 
0.03 
0.24 

3.0-4.2 
3-6 

2-7 
10-30 

2800-3300 
600-1300 
600-1000 
290-365 

250-400 
370-425 

3X1012 

1011 

4-5.5 

1-5.0 
1.9-3.2 
0.04 
0.24 
0.4-1.7 

I0n-1013 
109-1012 

1010-1012 

125 

125 

>1013 

>5X1018 

>8X101G 

>8XI01C 

>8X101G 

125 
250 
125 

500 
100 
600 

1013-I014 

'ixio14 

2.3 

3X1012 

ixio12 

8X109 

2.6-3.0 
2.5-4.0 
3.3-3.4 
2.5 
2.5 
6-8 
7-9 

0.25-0.6 

10" 

1012-1015 

3-3.7 
3.3-3.6 

5-7.5 
5-7.5 

0.5-2.5 
0.74-2.5 

0.7-4.0 
1.3-2.6 
5.0 
5.0 
1.0 
0.42 
0.82 
0.42 
0.3 
0.08 

10n 

125 

250-600 

2X1010 
2X1010 
>1013 
>1013 
2X1013 
1014 
5X1014 

3.2 
3.2 

3XI010 
3XI010 

5X109 
5X109 

3X107 
3X107 

<60 
>40 

>175 
>175 
1200 
860 

1.0 
0.45 

7 
7.6 

3.2 

8X1013 

5X1010 

2X106 
>1013 

4.8 

4.9 
4.7 
4.0 

0.2-0.4 

1300 
>500 
>500 
>180 
>180 

4.0 

0.05 
1.3 
0.15-0.2 
0.15-0.2 

3X1013 
I015 
1015 
1015-I017 

3X1012 
2.5X1015 

ixio10 

2.7X1014 
2X108 
2XI010 

8.3 

250 
250 

6.5-7.5 
7.8-8.3 

105-107 

1.4 

5X1012 

2X1012 

3XI08 

75 
200-500 
50-75 
320 

3.0-4.9 

1.8 

1.2 

3X1010 

ixio10 
ixio15 

5X109 

ixio11 

5X108 

6.1 

6.1 

0.18 

130 

1.5 

2X1010 
6X1011 

ixio9 
ixio8 

75-250 
<IOO 
<80 
235 
500-650 
300-700 

5.6 

0.3 

250 
250 

5.3 

1.0 

6.4 
41 

0.45 
37-32 

3.2X1011 

125 
8-16 

2.2 
8.3 

2.5-3.5 
9.4 
9.3 

0.25-0.6 
0.3-5.0 

0.2 
1.22 
0.52 

'2.9-6"" 
3.8-4.1 
1  1-13 

1X109 

ixio11 

2X106 
I011 
1012 

5XI011 
8X1010 

8X109 
3X109 
4X105 

ixio7 

2X107 

5.7-6.8 
8-9.5 
7.7 
6.9-7.5 
8-10 

"7-8.2" 
5.6-7.2 

4.2-5.3 
3.7-8 

8 

125 
125 
125 
125 

300 
270 
500 
300 

2XI08 

5.8-6.7 

11 

2.8-6 

I013 

2-28 


PKOPERTIES  OF  MATERIALS 


Physical  Properties  at  25°  C 


Table  1.    Properties  of  Solid 


Density, 
gm 
cm3 

Tens. 
Str., 
Ib 
in.2 

xio3 

Comp. 
Str., 
Ib 

in.2 

xio3 

Mod. 
of 
Elas- 
ticity 

xio6 

Flex. 
Str., 
Ib 

in.2 

xio3 

Coef. 
Lin. 
Ther. 
Exp. 
per  °C 

xio-6 

Ther. 
Cond. 
per°C 

xio~4 

(Note 
A) 

Max. 
Oper. 
Temp., 
°C 
(Note 
B) 

%H20 
Absorp- 
tion in 
24  Hr 

Material 
(See  tact  also) 
(Note  C) 

1.18-1.2 
1.18-1.2 
<1.2 
<1.2 

16-3.2 

2.6-3.2 
2.6-3.2 
16-3.2 
16-3.2 

2.3-2.4 

5.8-9 
6.5 
4-8 
5-10 

10-12.5 

0.3-0.5 

12-14 
>12 
12-16 
10-15 

70-90 

4-6 

62 
75 
50-58 
65? 

500 

500 
500 
500 
1000 

125 

<0.4 
<0.4 
0.4-0.6 
0.4-0.6 

Methacrylate,  cast,  regular  
"             "     heat-resistant. 
"           molded,  regular  .... 
"                "       heat-resist- 
ant 
Mica  (muscovite)  

10-15 
15-25 

0.4-0.6 
0.4 

60-80 
60-80 

5-7 
5-7 

12 

12 
12 
12 
12 

5-8 
5-8 

"     U.S.A.  clear  

"   India  clear  

"     stained  

"     (Phlogopite)  

"    reconstructed  plate  

"    flexible  

1.1 
40-50 
5-9 

Melt.  pt. 

Elongati< 

jn  650% 
1-1  6 

150 

Neoprene  GN  

i.06-U9 
1.06-1.19 
0.85-0.95 
0.8-1.0 
0.87-0.91 
0.86-0.88 

1.3-1.36 
1.3-1.36 
1.3-1.36 
1.3-1.36 
1.3-1.36 
13-1.36 
13-1.36 
13-1.36 
13-1.36 
13-1.36 
13-1.8 
1.5-1.8 
1.4-1.6 
1.2-1.3 
13-1.45 
135 
1.40-1.45 

1.6-10 

1.8-10 
1.27-132 
0.92 
138 
>1.05 
1.05-1.07 
13-1.4 
11-23 
23-23 
12-15 
I.I 
1.1 
I.I 
2.4 
12 
1.07 

1.1-1.4 
0.91 
0.92 

1.5? 

1.5? 

Nylon  *  filament  

18 
65-90°  C 

0.3-0.45 

12-15 

100 

120 

"       molded  

Ozokerite  

6.4 
4.7-6.2 
2  2-3.8 

Paper,  kraft,  dry  

Melt.  pt. 
Melt.  pt. 

9-15 
6-10 
8-12 
6-10 
6-8 
5-8 
7.5-12 
6.5-10 
7-11 
6.5-10 
5-12 
7-18 
14-19 
6-9 
6-8 
6-10 
6-7.5 

4-10 

>4.5 
5-12 
1.7-3.0 

58°  C 
38-50°  C 

35 
22 
34 
25 
32 
25 
35-44 
34-38 
30-38 
33-40 
30-40 
35^4 
42-47 
25-30 
20-30 
20-30 
25-30 

15-30 

15-30 
10-30 
3 

130-400 

Paraffin           

Petrolatum  

0.4-2.0 
0.4-2.0 
0.4-2.0 
0.4-10 
0.4-10 
0.4-2.0 
035-1.5 
035-1.5 
035-1.5 
035-1.5 
035-1.5 
035-1.5 
1-1.2 
0.7-1.0 
1-1.5 
1 
0.7-1.2 

1-1.5 

3.5? 
0.3-1.0 
0.015 

16-24 
11-19 
12-20 
12-20 
12-18 
12-18 
16-28 
13-21 
15-28 
15-28 
10-20 
16-30 
20-25 
11-17 
9-14 
11 
9-11 

8-20 

8-12 
9-15 
1.7 

17-25 
17-25 
17-25 
17-25 
17-25 
17-25 
17-30 
17-30 
17-30 
17-30 
17-25 
17-25 

5-8 
5-8 
5-8 
5-8 
5-8 
5-8 
5-8 
5-8 
5-8 
5-8 

100-110 
100-110 
100-110 
100-110 
100-110 
100-110 
100-125 
100-125 
100-125 
100-125 
150-200 
150-200 

4-6 
3-5 

1.3-2 
1.3-2 
1-1.2 
1-1.2 
2-4.4 
1.2-1.8 
1.5-2.5 
1.2-1.8 
1.0-1.5 
1.2-1.9 
0.3-0.5 
0.05-0.2 
0.5-1.6 
1 
1-1.75 

0.05-0.2 

0.007-0.07 
0.01-0.5 
NU-0.01 
0.05 
<0.1 
<0.1 
<0.1 
Nil 
Nil 
0.5-1.0 
High 

Phenolic  laminates,  Grade  X  

it                      (l                   It         T> 

"           "     XX.... 
"     XXP... 
"     XXX.. 
"     XXXP. 

"     CE  
"             "           "     L  
«     LE  
"             "           "     A 

"             "            "     AA  
"              "        glass  fabric  — 
"       moldings,  unfilled.  .  .  . 

25-60 
30-75 
30 
30-60 

25-40 

8-12 
4-7 
5 
4-7 

8-20 

125 
135-148 
135 
115 

150-200 

135 
70 
75-80 
82 
65 
65 
75 
200 
1000 
1000 
90 
90 
90 

1000  " 

"             "        wood  flour,  g.p. 
"             "        organic  elec.  .  .  , 
"             "          chopped    rag 
filled 
Phenolic  moldings,  heat-resistant 
asbestos  filled 
Phenolic  moldings,  mica  filled  
"       cast,  unfilled  

28 
170-210 

3-5 
6.2-8.1 

Polyethylene  

5-9 
>5 
>3 
2-4.5 
5-6 
1-2 

11.5-15 
14 

0.16-0.47 

>8-I2 
>8 
>8 
2 
10-11 
6-8 

60-80 
60-80 
60-80 
55 
3-6 
3-4 

1.9 
1.9 

Polystyrene,  general  purpose  
"          best  elcc 

1.7 
45-60 
30-50 

0.06? 
7-15 

5.8 
25-50 
25-50 
4.5 
4.5 
4.5 
21 
36 

Polytetrafluoroethylene  

Porcelain,  unglazed,  wet  proc  

"         oiled 

5 
7.0 
Melt.  pt. 

1.5-10 
0.13-035 
32-4.2 

45 
200.0 
70-1  00°  C 

2-5 

11 

""0.57* 

0.025 

Nil 

Prestite  * 

10.15 

•* 

033 

Quartz,  fused  

70-85 
670 
660 

3.8-8.7 

<65 

0.03-0.08 
2-4 
2-4 

Rubber,  hard  

3-3.8 

natural,  un  vulcanized  .... 
"            "      vulcanized  

"                  "          fin<K,  «i  nr*  r>vidr> 

*  Trademark  names. 

SOLID  DIELECTKIC  MATERIALS 


2-29 


Dielectric  Materials — Continued 


Electrical  Properties  at  25°  C 


Dielectric  Constant 
(Note  D) 

Power  Factor  % 
(Note  D) 

Volume  Resistivity 
(Note  E) 

Surface  Resistivity, 
ohms  at  20-25°  C 

Dielectric  Strength 
(Note  F) 

Refer- 
ence, 
See 
page  2-32 

Freq. 
less  than 
2kc 

300  to 
2000  kc 

Freq. 

less  than 
2kc 

300  to 
2000  kc 

Ohm-cm 
at 
20-25°  C 

Temp. 
Coef. 
20- 
30°  C 

Relative  Humidity 

Thick- 
ness, 
mils 

Volts 
per  mil 

30% 

50% 

90% 

<4 
<4 
<4 
<4 

4.5-7.5 

<3 
<3 
<3 
<3 

6-7 
<7 
3.8-7 
7 

0.1-7.0 

1.5-4.0 
<4 
1.5-4 
<4 

>1014 
>1014 
>1014 
>1014 

1.3X10U- 
2X1017 

>1012 

125 

500 

40 
40 
40 

40 

1,7,12,16 

14 
14 
1,14 

12,14 

39 
39 
40 
39 
32,40 
12 
12 
1,5,12 
10,12 

40,41 
40,41 
40,41 
40,41 
40,41 
40,41 
40,41 
40,41 
40,41 
40,41 
40,41 
40,41 
39 
40 
40 
40 
40 

40 

40 
40 
39 
33 

40 
40 
40 
39 
1,  12,  13 
19,39 
12 
11 
11 
39 
1,9,  13 
1,8 

1,  4,  5,  12 
6 
6 
i 

125 

500 

10 

ixio14 

2X1013 

8X109 

6.57-8.59 
7.07-7.90 
5.83-9.64 
5.41-6.07 

3.4-4.1 

0.01-0.04 

1-11 
1-11 
1-11 
1-11 

6 
6 

1500-5700 
1300-4200 
1300-4100 
1500-5000 

950 
600 

0.01-0.02 
0.06-8.36 
0.38-7.12 

2.7 

I'Jxio1* 

Txib1" 

'9X107 

4.5X1011- 

0.13-0.32 

2X1013 
2X1011 

8.3 
4.5 
3.2-4.5 
2.4 
3.5 
2.2 
2.2 

1.6 
2.7 
1.0-2 
0.92 
0.5 

4.2X1012 
10U 
1013 
5X10U 

2.2-2.5 

3.3-4 

2X1010 

125 
25 
6 

400 
1100 
750-1000 

2.2 

0.03 
0.29-0.5 

0.02 

>5X1018 
2X1012- 
1013 

>1X1018 

>ixio18 

100 

62 
62 
62 
62 
62 
62 
62 
62 
62 
62 
125 
125 
62 
125 
125 
125 
125 

125 

125 
125 
62 

500 

500-700 
500-700 
500-700 
500-700 
500-650 
500-650 
200 
400-500 
200 
400-500 
110-225 
50-150 
500-600 
300 
200-350 
550 
400 

300-600 

550 
350-450 
1000-1100 

4.7-5.5 
4.7-5.5 

3.8-4.5 
3.8-4.5 

,0io_,0i3 

5.6XI09 

4.5-5.2 
4.2-5.2 
7 
5-6 
7 
4.5-5.5 

5.0 

3.0-3.5 
2.4-3.0 
10 
4.5-6.5 
10 
3.5-5.5 

'  Yo9-io12' 

I09-1012 
I09-1012 
109-1012 

3XI09 
9X108 

12.0 

1.8X109 

7.5 

15 

0.9-1.3 

3.7-4.1 

5-7 
5-12 
6.25 
5-10 

5-20 

6  max. 
5-10 
2.3 

4.5-6 
4.5-8 
5.5 
4.5-6.5 

4.5-20 

4.5-5 
5-7 
2.3 
2.6 
2.5-2.75 
2.5-2.7 
2.6-2.8 
2.0 
6-7 
6-7.5 

5-10 
4-35 
7.3 
8-20 

10-18 

1.0-2.5 
2.5-20 
0.03-0.05 

1.5-3.0 
3.5-9 
4.3 
5-10 

4-10 

0.7-1.5 
1.0-4.5 
0.03-0.05 
0.02 

1012 
108 
1011 
109-10n 

109-10n 

1012max. 
109-1014 
1016-1017 

2.4X108 
I.6X1010 
108-1010 

1014 

1.7X1010 
IQiQ-io12 
2.4X1  010 
>I013 

2.5-2.75 
2.5-2.7 
2.6-2.8 
2.0 
6-7 
6-7.5 
2.9-4.5 
5.0 
3.0 
7.65 

0.01-0.05 
0.01-0.04 
0.04-0.3 
<0.02 
1.0-2.5 
1.7-2.5 

0.02-0.1 
0.016-0.04 
0.02-0.1 
<0.02 
0.8-1.0 
0.8-1.0 

1016 
1017-1019 
3.5X1019 
1016 
1014 
108-1012 

ixio9 

125 
125 
125 
80 
250 
250 
80-120 
60 
60 

>450 
500-700 
400 
480 
250 
40-100 
125-300 
750 
400 

>1017 

>1013 
5X106 

1.6 

2X1013 

6X1011 

6.08 
4.2 

2.76 

0.9 
0.02 
0.25-0.4 

0.6-2.1 
0.1-0.2 
0.4 

6X1011 
>5X1018 
5X1015- 
5X1016 
1018 
101S-1015 
1015-1015 
3.5X1015 

"*3.6" 

ixio15 

8X1014 
6X1015 

3X1012 
5X1014 

3X1015 

2X108 
2X1014 

2X109 

500 

200 

2.73 

2.8-3.5 
2.3-2.5 
2.4-2.9 
5.01 

3.3-4.7 

3 
2.3-2.4 
2.4-2.7 

0.287 

0.4-0.5 
0.1-0.3 
0.4 
0.81 

80 

250-900 

500-700 

2-30 


PROPERTIES  OF  MATERIALS 


Physical  Properties  at  25°  C 


Table  1.    Properties  of  Solid 


Density, 
gm 
cm3 

Tens. 
Str., 
Ib 
in.2 

xio3 

Comp. 

Str., 
Ib 
in.2 

xio3 

Mod. 
of 
Elas- 
ticity 

xio6 

Flex. 
Str., 
Ib 
in.2 

xio3 

Coef. 
Lin. 
Ther. 
Exp. 
per  °C 
X10"G 

Ther. 
Cond. 

per°C 

xio~4 

(Note 
A) 

Max. 
Oper. 
Temp., 
°C 
(Note 
B) 

%H20 

Absorp- 
tion in 
24  Hr 

Material 
(See  text  also) 
(Note  C) 

Rubber,  natural,  20%  carbon'black 
"     wire  insulation,  Code  
Perform- 
ance 
Rubber,     wire    insulation,    heat- 
resistant 
Rubber,  wire  insulation,  low  water 
absorption 
Rubber,  wire  insulation,  low  power 
factor 
Rubber,  synthetic,  Buna  S,  gum 
stock 
Rubber,  synthetic,  Buna  N,  gum 
stock 
Rubber,    synthetic,    butyl,    gum 
stock 
Rubber,  synthetic,'polyisobutylene, 
gum  stock 
Rubber,  synthetic,  Neoprene,  gum 
stock 
Rubber,  synthetic,  polysulfide,  gum 
stock 
Rubber,  synthetic,  cyclicized  natu- 
ral rubber 
Saran,*  molded  

>0.5 
>1.2 

>1.5 
>1.5 
>1.5 
1.5-3.2 
2-4.5 
3.2 
0.3-1.5 
1.6-1.75 
0.8-1.4 

Elongation  >200% 
Elongation  >400% 

Elongation  >400% 
Elongation  >450% 
Elongation  >450% 
Elongation  400-650% 
Elongation  400-800% 
Elongation  800% 
Elongation  600-1  000% 
Elongation  40M35% 
Elongation  350-600% 

50 
60 

75 
70 
70 
120-140 
120-140 
145 
60 
145 
90 
<50 
70-90 



j 



0.94-0.98 
0.96-1.03 
0.91 
0.91 
1.24 
1.34-1.6 
0.97 

1.65-1.75 
1.009 
1.1-2.7 

<0.1 

00-0.1 

4-6 
Melt.  pt. 
0.9-2.0 

7.5-10 

45-75°  C 
7 

0.04-0.24 

15-17 

160-190 

2.2 
6.0 

Shellac  

3.0 

40-60 

150-200 
175 
175 
200? 

<0.25 
<03 

"      compound  

28-30 

Silicone  glass  laminate  

"      varnish  

"      varnished  glass  cloth  
"      sealing  compound  
"      fluid 

0.98-1.00 
0.968-0.973 
1.5-2,0 
2.6-33 
2.5-2.6 
2.5-2.8 
0.95-0.97 
136 
1.38 
2.0-2.1 

0.20-0.33 
3.5-10 
6.5-10 
6.5-10 
0.9-1.2 
33 

"      rubber  (Silastic  *) 

10-14.2 
65-90 
65-90 

8 
13-15 
13-15 

>8 
18-24 
18-24 

10.5-20 
6.5-8.5 
6.5-8.5 
180-220 

48 
60 
60 
432 

120-200 
980-1000 
980-1000 
60-90 
80 
110 
95 
200 

300 

0.2 
Nil-0.1 
Nil-0.1 
0.2-0.5 
0.05 
0.03 

Slate  

Steatite  ceramics,  regular.  . 

"            "       low  loss  .  . 

Styraloy  *  22 

11 

033 

6.5 

Styramic  *  

"          HT 

Melt.  pt. 
1.2 

5-7.5 

1  12.8°  C 
1.14 

80-100 

64 

7 

Sulfur.  ... 

Tegit  * 

3.9-4.05 

15 

18-20 

7-8 

Nil 

Titanium  ceramics: 

Barium  titanate 

4 

3-3.6 

Calcium  titanate 

4-5 

40-80 

16-19 

5-7 

300 

Nil 

Magnesium  titanate 

1.45-1.6 
<1.55 

6-10 
>6 

24-30 
>24 

1.2-1.5 

>IO 
>9 

25-30 

7 

77 
77 

1-3 

1-3 

Urea  formaldehyde,  cellulose  filled. 
"             "          arc  resistant  .  . 

1.24 
1.26 

U6 
1.05-13 

Tens,  sta 
Tens,  sta 

2-6 

1-7 

.401b.pe 
.  40Ib.pt 

r  in.  widt 
riru  widt 

i 

ii 

5.0 
6.0 

5.0 

85 
90 

100 
60 

Varnished  cloth  yellow 

Attacked 
<3.0 

"           "     black 

Vinyl  plastics: 
Polyvinyl  alcohol 

150 

«       butyral 

3-14 
1-9 

>8 
>6 

65 

ca.4.0 
4.0 

70-100 
60-80 

45 

47 

13 
0.4-1.0 

<0.15 
<0.15 

u       caroazoie  

1.22-1.65 

132-136 
1.4-1.55 

'       chloride,  filled,  non-rigid 
Vinyl  chloride-acetate: 
Clear  sheets  

>038 
>036 

>13 
>11 

69 

Sheets  

*  Trademark  names. 

SOLID  DIELECTRIC  MATERIALS 


2-31 


Dielectric  Materials — Continued 


Electrical  Properties  at  25°  C 


Dielectric  Constant 
(Note  D) 

Power  Factor  % 
(Note  D) 

Volume  Resistivity 
(Note  E) 

Surface  Resistivity, 
ohms  at  20-25°  C 

Dielectric  Strength 
(Note  F) 

Refer- 
ence, 
See 
page  2-32 

Freq. 
leas  than 
2kc 

300  to 
2000  kc 

Freq. 
less  than 
2kc 

300  to 
2000  kc 

OhnMm 
at 
20-25°  C 

Temp. 
Coef. 
20- 
30°  C 

Relative  Humidity 

Thick- 
ness, 
mils 

Volts 
per  mil 

30% 

50% 

90% 

5.97 
4.5-6.0 
5.0-6.0 

5.0-6.0 
2.75-3.0 
2.75-3.0 
2.7-4.4 
14-19 
2.1 
2.2-2.4 
7.5 
7.5 

8.8 

3X1013 
>5X1014 
>2.7X1015 

>2.0X1015 
>3.2X1015 
>2.7X1015 

6 

28,39 
28,39 

28,39 
39 
39 
40 
40 
40 
40 
40 
40 
40 

28,39 
1,7,8,12 

40 

39 
39 
39 
39 
39 
39 
1,5,6,12 
39 
39 
39 
40 
33 
1,16 
1,12 

37,39 
37 
37 
37 
37 
40 
40 
5 
2,  3,  II! 
2,3,11 

40 
40 
40 
40 
40 

40 
40 

5.0-7.0 
4.0-6.0 

2.4 
2.4 

2.4 

375-425 
450-550 

450-550 
600-700 

4.0-6.0 
0.8-1.3 
0.8-1.3 

1.7 

1015-1016 
109-10n 
1018 
1016-1018 

750 
500 

3.8-9.0 
004 

0.02-0.05 

3 
50 

400-500 

2.6-2.7 

3-3.3 

4.1 

0.06-0.12 

4.5-6.5 
2.5 

1016 

1014-1015 
IXIO16 
6X1010- 
2.3XI011 

620 

>350 
900 
200 

4-6 
S-3.7 

3-8 
0.81 

125 
0.8 
200 

1.5 

2X1014 

5xib13 
ixio12 

6X109 

<0.5 

3-3.5 

3-4 
2.8 
2.4-2.85 

0.7 
0.3-0.7 
0.05-0.07 
0.01 

7 
10 
100 

1500-2000 
500 
250-300 
500 
5-10 
200-250 
200-250 
700-800 
800 

2.8 
Z4-2.82 
5-7.5 
<30 
5.5-6.5 
5.5-6.5 
2.4-2.6 
2.5 
2.6 
3.8 

0.05 
0.02-0.06 
013-018 

1013-1014 
,0H 

>1013 

ixio6 

>I09 
>6XI08 

6-7.5 
5.5-6.5 
5.5-6.5 
2.5-2.6 
2.55 

8.6 
0.13-0.3 
<0.15 
0.07-0.12 
0.11-0.26 

<63 
0.1-0.2 
0.04-0.1 
0.05-0.15 
0.04 
0.02 

ixio8 

1014-1016 
1014-I016 

,020 

2X1  0s 

9X106 

1000 
250 
250 
125 

1010-1012 

uxio16 

5X107 

3.6-4.22 

ixio17 

2X1012 

JQlS-IO" 

4.9 
1.4 

ixio16 

2X1012 

7X1015 

50 

100-200 

80-100 
1200 
165 
14-18 
275 
6.6-8.2 
<8.2 
48 

0.04-0.07 
<0.1 
<01 

250 

0.007 
<0.1 

>1014 

250 

100-200 

7.5-9.5 
7.5-9.5 

4-6 
<10 

2.7-4.6 
<4.0 
512 

10n-1013 
>1010 

10n-1012 

125 
125 
1 
10 
10 

300-720 
300-720 
700-1000 
900-1200 
800-1100 

8X1013 

ixio13 

ixio9 

4.5-5.5 
4.5-5.5 

2.5 
2.0 

8 
6 

3.0 
2.0 

109 

6 

107 
10" 

3.6-3.7 

3.3-3.5 

2.9 
3.7 

6 

300-400 

0.4 

0.9 

3.8 
4-12 

3.2-3.5 
3.2-3.5 

0.7 
13.6 

<1.3 

<U 

1016 

500-700 
600-2000 

600 
400 

75 

125 

3-3.3 
3-3.3 

<1.9 
<1.9 

>1014 
>1014 

4X1010 

2-32 


PROPERTIES  OF  MATERIALS 


Table  1.    Properties  of  Solid 


Physical  Properties  at  25°  C 


Density, 
gm 
cm3 

Tens. 
Str., 
Ib 
in.2 

xio3 

Comp. 

Str., 
Ib 
in.2 

xio3 

Mod. 
of 

Elas- 
ticity 

xio5 

Flex. 
Str., 
Ib 
in.2 

xio3 

Coef. 
Lin. 
Ther. 

Exp. 
per°C 

xio~6 

Ther. 
Cond. 
per°C 

xio~4 

(Note 
A) 

Max. 
Oper. 
Temp., 
°C 
(Note 
B) 

%H20 
Absorp- 
tion in 
24  Hr 

Material 
(See  text  also) 
(Note  C) 

•s.7  S 

Q_  17 

n  ^^_A  A 

i?  n 

69 

4 

45 

<0  15 

Vinyl  chloride-acetate  (Contd.) 
Transparent,  molded  

1  3-1  4 

v,4 

^.q 

75-12 

45 

<0.15 

Molded  

i  21  4 

v,C 

9-12 

035-085 

8-12 

47 

<0.!5 

Opaque  molded  

1  15-1  29 

1  3 

60-70 

0.3-0.7 

Non-rigid  

1  30-1  45 

1730 

60-70 

0.5-0.9 

Filled,  non-rigid  

T    £O1    "Jt 

X    A 

0_1fl 

flflft-fl  17 

ic_j7 

190 

2  2 

70-90 

0  1 

Vinylidene  chloride  

0  62-0  75 

0  77 

6-86 

70 

64 

43-104 

Wood,  maple,  hard  

"        "     paraffined  

ft  AQ-fl  Oft 

0  77 

6-72 

130 

49 

5-8 

"     oak  

3.7 

12.7 

90 

25 

4.9 

120 

1000 

Nil 

Zircon  porcelain  

Note  A.  Thermal  conductivity  is  expressed  in  10  4  X  gram-calories  per  square  centimeter  per 
second  for  a  temperature  gradient  of  1  deg  cent  per  centimeter.  The  values  are  typical  for  the  tempera- 
ture range  0  to  100  deg  cent.  The  temperature  gradient  is  perpendicular  to  the  laminations  of  laminar 
materials. 

Note  B.  Maximum  operating  temperatures  given  are  based  on  satisfactory  operation  under  average 
conditions  without  excessive  cold  flow,  distortion,  or  shortening  of  the  operating  life  of  the  material 
In  many  cases  it  will  be  necessary  to  reduce  the  operating  temperature  in  order  to  obtain  electrical 
properties  or  to  reduce  cold  flow  at  higher  unit  stresses.  In  some  cases  higher  operating  temperatures 
may  be  used,  particularly  where  the  material  can  be  obtained  in  special  grades  for  this  purpose.  If 
thermal  shock  is  involved  the  limit  for  many  materials,  especially  the  ceramics,  will  be  much  reduced. 

Note  C.  Most  of  the  materials  listed  are  intended  to  be  representative  of  a  class  rather  than  of  a 
single  sample,  and  the  ranges  of  values  should  be  interpreted  accordingly.  Since  complete  mechanical 
and  electrical  data  seldom  are  available  for  a  single  sample  or  batch,  and  since  data  have  been  assem- 
bled from  many  sources,  it  is  necessary  to  exercise  good  judgment  in  comparing  the  various  materials. 
Values  for  laminated  phenolic  materials  are  those  for  sheets;  properties  of  tubes  are  somewhat 
poorer. 

Note  D.  Where  possible,  the  ranges  of  dielectric  constant  and  power  factor  include  the  variations 
to  be  expected  over  the  indicated  frequency  ranges,  although  most  of  the  values  up  to  2  kc  are  for  60 


REFERENCES 

1.  Curtis,  H.  L.,  Insulating  Properties  of  Solid  Dielectrics,  Sci.  Papers  Bur.  Standards,  No.  234 
*     (1915). 

2.  Taylor,  T.  S.,  The  Thermal  Conductivity  of  Insulating  and  Other  Materials,  Elec.  /.,  December 

1919,  pp.  526-532. 

3.  Flight,  W.  S.,  /.  I.E.E.,  Vol.  60,  218-235  (1922). 

4.  Hoch,  E.  T.,  Power  Losses  in  Insulating  Materials,  Bell  Sys.  Tech.  /.,  November  1922. 

5.  Preston,  J.  L.,  and  E.  L.  Hall,  Radio  Frequency  Properties  of  Insulating  Materials,  QST,  Vol.  9 

26-28  (February  1925). 

6.  Curtis,  H.  L.,  and  A.  T.  McPherson,  Dielectric  Constant,  Power  Factor  and  Resistivity  of  Rubber 

and  Gutta  Percha,  Technologic  Papers  Bur.  Standards,  Vol.  19,  669-722  (May  1925). 

7.  Monkhouse,  A.T  Electrical  Insulating  Materials.    Sir  Isaac  Pitman  and  Sons,  London  (1926). 

8.  Lee,  J.  A.,  and  H.  H.  Lowry,  Ind.  Eng.  Chem.,  February  1927,  pp.  387-395. 

9.  Sosman,  R.  B.,  The  Properties  of  Silica.    Chemical  Catalog  Co.,  New  York  (1923). 

10.  Shanklin,  G.  B.,  and  G.  M.  J.  Mackay,  Progress  in  High  Tension  Underground  Cable  Research 

and  Development,  Trans.  Am.  Inst.  Elec.  Eng.,  Vol.  48,  338-372  (1929). 

11.  Peek,  F.  W,,  Jr.,  Dielectric  Phenomena  in  High  Voltage  Engineering.    McGraw-Hill    New  York 

(1929). 

12.  International  Critical  Tables.    McGraw-Hill,  New  York  (1929). 

13.  Barringer,  L.  E.,  Mycalex— A  Moulding  Material  with  Unique  Properties,  Gen  Eke   Rev    Julv 

1931,  pp.  406-409.  ' 

14.  Lewis,  A.  B.,  E.  L.  Hall  and  F.  R.  CaldweU,  Some  Electrical  Properties  of  Foreign  and  Domestic 

Micas  and  the  Effect  of  Elevated  Temperature  on  Micas,  Bur.  Standards  J.  Research,  August 
1931,  pp.  403-418. 

15.  Littleton,  J.  T.,  and  G.  W.  Morey,  Electrical  Properties  of  Glass.    John  Wiley  New  York  (1933) 

16.  Smithsonian  Physical  Tables  (1933). 

17.  Abraham,  H.,  Asphalts  and  AUied  Substances.    Van  Nostrand,  New  York  (1938). 

18.  Hartshorn,  L.,  Plastics  as  Insulators,  J.  I.E.E.,  Vol.  38,  474  (1938). 

19.  Thurnauer,  H.,  Ceramic  Insulating  Materials,  Elec.  Eng.,  Vol.  59,  451  (November  1940) 

^230  481  (1940^  GIa3SOW'  CoiQI)ressed  Magnesia  as  an  Electrical  Insulator,  /.  Franklin  Inst., 


SOLID  DIELECTRIC  MATERIALS 


2-33 


Dielectric  Materials  —  Concluded 


Electrical  Properties  at  25°  C 


Dielectric  Constant 
(Note  D) 

Power  Factor  % 
(Note  D) 

Volume  Resistivity 
(Note  E) 

Surface  Resistivity, 
ohms  at  20-25°  C 

Dielectric  Strength 
(Note  F) 

Refer- 
ence, 
See 
page  2-32 

Freq. 
less  than 
2kc 

300  to 
2000  kc 

Freq. 
less  than 
2kc 

300  to 
2000  kc 

Ohm-cm 
at 
20-25°  C 

Temp.. 
Coef. 
20- 
30°  C 

Relative  Humidity 

Thick- 
ness, 
mils 

Volts 
per  mil 

30% 

50% 

90% 

3.2-3.5 

3-4.0 

1-4 

1.8 

>10U 
>1014 
>10U 
>108 
104-108 
>1014 

125 
125 

>600 
>650 

40 
40 
40 
40 
40 
40 
2,  3,  16] 
1,12 
3,16 
36 

4.7 

4 

4-4.8 

1-3 

75 
75 
125 

>400 
>400 
350-2000 

6-9.5 
3-5.1 

6-12 
3-8 

3-4 
4.4 

3-6.5 
3.33 

4.1 
3.64-6.84 

3XI010 

3.6 

1XI012 

8X10U 

2X109 

600 

110 

3.3 
9.2 

3.85 
0.13 

ca.  IOU 

250 

240 

cycles  or  1  kc,  and  most  of  the  values  in  the  higher  frequency  ranges  are  for  1  Me.  For  good  insulating 
materials,  the  dielectric  constant  does  not  change  greatly  at  higher  frequencies,  but  the  power  factor 
may  rise  or  fall  considerably.  An  auxiliary  table  of  power  factors,  following  the  main  table,  gives  data 
for  some  substances. 

Note  E.  Volume  resistivity  has  a  large  negative  temperature  coefficient.  The  values  shown  for 
temperature  coefficient  are  the  ratio  of  the  resistivity  at  20  deg  cent  to  the  resistivity  at  30  deg  cent. 
It  should  be  remembered  that  at  higher  operating  temperatures  the  order  of  merit  of  any  two  materials 
may  be  reversed.  Furthermore,  the  volume  resistivity  may  be  seriously  reduced  by  prolonged  exposure 
to  high  humidities. 

Note  F.  Dielectric  strength  given  is  for  the  short-time  test  at  60  cycles  except  in  the  case  of  the 
ceramics,  where  it  is  given  for  the  step-by-step  method.  Only  those  values  for  the  same  thickness  are 
directly  comparable.  Where  two  thicknesses  are  given,  the  higher  dielectric  strength  applies  to  the 
lower  thickness.  The  dielectric  strength  will  be  much  less  at  higher  temperatures  or  for  long  times 
(see  discussion  of  dielectric  strength) .  In  the  case  of  laminated  materials,  the  electric  field  is  perpendicu- 
lar to  the  laminations;  dielectric  strength  with  the  field  parallel  to  the  laminations  may  be  very  low. 

Note  G.  For  all  wood  the  tensile  strength  and  flexural  strength  are  given  for  forces  perpendicular 
to  the  grain,  the  compressive  strength  for  forces  parallel  to  the  grain.  Under  Thermal  Conductivity, 
the  first  figure  is  perpendicular,  the  second  parallel,  to  the  grain. 

21.  Rosenthal,  E.,  The  Electrical  Properties  of  High  Frequency  Ceramics,  Electronic  Eng.,  September 

and  October  1941. 

22.  Rosenthal,  E.,  and  J.  E.  Nickless,  Ceramic  High  Frequency  Insulators,  Wireless  World,  Novem- 

ber 1941. 

23.  Shand,  E.  B.,  The  Dielectric  Strength  of  Glass — an  Engineering  Viewpoint,  Elec.  Eng.,  Vol.  60,  41 

(March  1941). 

24.  Simons,  H.  R.,  and  C.  Ellis,  Handbook  of  Plastics.    Van  Nostrand,  New  York  (1943). 

25.  Technical  Data  on  Plastics  Materials.    Plastics  Materials  Manufacturers'  Association,  Washington, 

D.  C.  (1943). 

26.  Scribner,  G.  K.,  A  Ready  Reference  for  Plastics.    Boonton  Molding  Co.,  Boonton,  N.  J.  (1944). 

27.  Guyer,  E.  M.,  Electrical  Glass,  Proc.  I.R.E.,  Vol.  32,  743  (December  1944). 

28.  ASTM  Standards,  Vol.  Ill  (1944)  (specifications  for  various  plastics). 

29.  Englund,  C.  RM  Dielectric  Constants  and  Power  Factors  at  Centimeter  Wave  Lengths,  Bdl  Sys. 

Tech.  J.,  Vol.  23,  114  (January  1944). 

30.  Works,  C.  N.,  T.  W.  Daldn  and  F.  W.  Boggs.    A  Resonant-Cavity  Method  for  Measuring  Dielec- 

tric Properties  at  Ultra  High  Frequencies,  Proc.  I.R.E.,  Vol.  33,  245  (April  1945). 

31.  Barren,  H.,  Modern  Plastics.    John  Wiley,  New  York  (1945). 

32.  Field,  R.  F.,  The  Effect  of  Humidity  on  Plastics  Insulation,  Plastics  and  Resins,  September  1945. 

33.  Brouse,  H.  L.,  A  Review  of  Plastic  Materials,  Proc.  I.R.E.,  Vol.  33,  825  (December  1945). 

34.  Bass,  S.  L.,  and  T.  A.  Kauppi,  Silicones— a  New  Class  of  High  Polymers  of  Interest  to  the  Radio 

Industry,  Proc.  I.R.E.,  Vol.  33,  441  (July  1945). 

35.  Plastics  Design  is  Given  Impetus  by  Engineering  Classification  (SPI),  Elec.  Mfg.,  Vol.  36,  128 

(November  1945). 

36.  Russell,  R.,  and  W.  C.  Mohr,  Zircon  Porcelain,  New  Insulation  for  New  Products,  Elec.  Mfg., 

January  1946. 

37.  Wainer,  E.,  High  Titania  Dielectrics,  Electrochem.  Soc.  Trans.,  1946. 

38.  Warner,  A.  J.,  Problems  in  the  Manufacture  of  Ultra-high-frequency  Solid-dielectric  Cables,  Proc. 

I.R.E.,  Vol.  1  of  Waves  and  Electrons  section,  p.  31W  (January  1946). 

39.  Manufacturers'  catalogs,  circulars,  and  other  data. 

40.  References  24,  25,  26,  28,  31,  33,  35,  39. 

41.  NEM A  Standards. 


2-34 


PROPERTIES  OF  MATERIALS 


Auxiliary  Table.    Power  Factor  of  Insulating  Materials  at  High  Frequencies 

(Approximate  representative  values) 


Material 

Frequency 

60  cycles 

1  kc 

1  Me 

10  Me 

100  Me 

1000 

Me 

10,000 

Me 

0.0002 
.0002 
.00024 
.0003 
.0002 
.0005 

0.0002 
.0002 
.0002 
.0002 
.0002 
.0003 
.0004 
.0004 
.0005 

0.0002 
.0002 
.0002 
.0002 
.0002 
.0003 
.00035 

0.0002 
.0002 
.0002 
.0002 
.0003 
.0003 
.00035 

0.0001 
.0002 
.0002 
.0002 
.0004 
.0004 

0.0001 
.004 

Polytetrafluoroethylene     .... 

0.0002 
.00024 

.0003 
.0004 
.0004 
.0014 
.0005 
.001 

.0002 
.0005 
.0014 
.0004 

.0004 

.0005 
.0008 

.0008 
.0008 
.0009 
.001 
.0015 
.0034 
.004 
.005 
.006 

VJ             ,    MI£M 

.06 

.0026 

.0013 
.004 

.0008 

.0009 
.003 
.002 
.0035 
.004 
.006 
.006 
.015 
.01 

Polyvinyl  carbazo  e  

.003 
.0021 
.003 
.004 
.005 
.0033 
.006 
.014 
.02 
.04 

.0014 
.002 
.003 
.006 
.004 
.006 
.005 
.007 
.0086 
.03 
.015 

.0011 
.0015 
.003 
.005 
.004 

Steatite  low  loss     

.0022 
.015 
.003 
.006 

.0025 

\niline  formaldehyde 

.008 

Glass  Pyrex 

Rubber  hard  (best) 

Ethvl  cfllulos?  (best) 

.007 
.025 
.03 
.03 
.015 

.006 

.025 

Phenolic,  mica  filled  (best)  .  .  . 

.010 

.01 

orce    in,     e   p  oc 

.03 

Vinyl  resin  (hard) 

.007 

Phenolic  glass  base     

.011 
.035 

.013 
.04 

.018 
.05 

.02 
.080 

.024 
.04 

Phenolic,  paper  base  

.05 

.04 

ADDITIONAL  INFORMATION  ON  SOLID  DIELECTRIC  MATERIALS. 

Acrylic  Resins  and  Acrylates.  Designations  used  for  thermoplastic  polymers  derived  from  acrylic 
acid  (CH3  :  CHCOOH),  methacrylic  acid  [CH2  :  C(CH3)COOH],  or  allied  materials.  The  most  impor- 
tant are  the  methyl  methacrylates,  q.v.  Acrylics  may  be  vulcanized  to  reduce  thermoplasticity. 

Alkyd  Resins.  Resins  derived  from  reaction  of  polybasic  acids  with  polybasic  alcohols,  glycerol 
and  phthalic  anhydride  being  the  principal  materials  in  use.  They  are  used  chiefly  for  varnishes  and 
other  coatings.  Trade  name:  Glyptal. 

Allyl  Resins.  A  group  of  plastics  derived  from  allyl  alcohol  (CH2  :  CHCHaOH).  The  resins  are 
marketed  in  the  form  of  cast  transparent  sheets  and  rods,  and  in  the  form  of  liquid  monomers  which 
are  polymerized  with  the  aid  of  a  peroxide  catalyst  and  heat,  but  no  gases  or  vapors  are  emitted.  The 
resin  is  thermosetting  in  nature  with  good  electrical  properties  which  are  maintained  at  elevated  tem- 
peratures. Index  of  refraction,  UD,  is  1.49  to  1.51.  Light  transmission  is  over  91  per  cent.  Specifica- 
tions: ASTM  D819.  Trade  names:  Allymer,  MR  Resins,  Kriston. 

Allymer  *  CR-39  et  al.     Pittsburgh  Plate  Glass  Co.    Allyl  resins. 

Alsimag.*     American  Lava  Corp.     Steatite,  cordierite,  and  other  ceramic  bodies. 

Alvar.*    Shawinigan  Products  Corp.    Polyvinyl  acetal  resin. 

Amber.  Yellow  or  orange  fossil  resin  found  on  the  shore  of  the  Baltic  Sea.  High  insulation  re- 
sistance makes  it  an  excellent  insulator  for  electrometers. 

Amphenol  *  912.     American  Phenolic  Corp.    Polystyrene  sheet,  rod,  and  tubes. 

Amphenol  *  912B.     American  Phenolic  Corp.     Methacrylate  sheet,  rod,  and  tubes. 

Aniline-formaldehyde  Resins.  Derived  from  aniline  and  formaldehyde  and  fabricated  by  casting- 
and  very  limited  molding  under  heat  and  pressure.  The  resin  is  but  slightly  thermoplastic,  and  parts 
are  usually  made  by  machining  sheet  or  rod  stock.  Color  is  reddish  brown.  Insulation  resistance  is 
high  and  dielectric  losses  are  low  over  a  wide  frequency  band.  Trade  names:  Dilectene,  Cibanite. 

Annite.*     Spaulding  Fibre  Co.     Thin  vulcanized  fiber  (fish  paper). 

Aroclors.*     Monsanto  Chemical  Co.     Chlorinated  diphenyl  resins  and  oils. 

Asbestos,  A  hydrated  magnesium  silicate  mineral  in  fibrous  form.  Two  distinct  groups  of  min- 
erals are  described  as  asbestos:  amphibole,  or  hornblende  asbestos,  in  various  subgroups,  most  of  which 
contain  iron  and  have  harsh  and  springy  fibers,  relatively  weak  and  non-flexible;  and  serpentine  as- 
bestos, the  principal  subgroup  consisting  of  chrysotile,  so-called  Canadian  asbestos,  which  has  soft* 
fine,  strong  fibers,  suitable  for  manufacture  of  asbestos  textiles.  Chrysotile  asbestos  is  stable  up  to. 
temperatures  of  400^ to  500  deg  cent  and  may  be  useful  at  still  higher  temperatures.  It  is  not  a  par- 
ticularly good  electrical  insulator  and  has  very  high  losses  when  moist  but  is  valued  chiefly  for  its  fire 
resistance  and  its  heat  resistance.  Asbestos  products  should  not  be  allowed  to  come  into  contact  with 
fine  wires  since  corrosion  may  result. 

*  Trademark  names. 


SOLID  DIELECTRIC   MATERIALS  2-35 

Asbestos  Paper.  Made  by  felting  asbestos  fibers,  often  with  some  rag  fibers  for  additional  strength. 
It  is  employed  normally  only  as  a  flame  barrier  or  for  heat  insulation  and  should  not  be  depended  upon 
for  electrical  insulation. 

Asbestos  Textiles.  May  contain  up  to  6  per  cent  total  iron  content  and  as  much  as  2  per  cent 
iron  in  a  magnetic  form.  A  higher  grade  known  as  "non-ferrous,"  containing  less  than  1.75  per  cent 
total  iron  and  less  than  0.75  per  cent  magnetic  iron,  is  available.  Non-impregnated  asbestos  products 
are  extremely  hygroscopic.  In  order  to  avoid  absorption  of  moisture,  the  products  often  are  treated 
with  varnishes  or  oils,  but  impregnation  reduces  the  maximum  operating  temperature  sharply. 

Asbestos  Wood.  Asbestos  combined  with  Portland  cement  to  form  dense  sheets  (Transite).  Also 
combined  with  magnesia  and  cement  in  an  insulating  board  impregnated  with  a  black  insulating 
compound  (ebony  asbestos  wood). 

Asbestos  Ebony.*  Johns-Manville  Co.  Asbestos  fiber  and  binding  cement  impregnated  with 
compound. 

Asphalts,  Natural.  Native  asphalts  are  roughly  divided  into  relatively  pure  deposits  containing 
less  than  10  per  cent  mineral  matter  and  those  containing  a  large  amount  of  mineral  matter.  Both 
types  of  deposit  contain  water,  but  in  the  latter  group  the  water  is  often  in  emulsified  form.  The 
water  content  may  be  as  high  as  40  per  cent.  It  is  very  difficult  to  separate  any  but  the  largest  particles 
of  mineral  matter  by  heating.  Asphalt  is  used  for  potting  and  impregnating  compounds,  in  the  manu- 
facture of  varnishes  and  japans,  and  for  the  insulating  covering  of  cables.  It  is  closely  related  to  petro- 
leum asphalt  and  asphaltites,  which  are  used  for  the  same  purposes.  Refined  Trinidad  asphalt  melts 
at  87  deg  cent  and  contains  about  38  per  cent  mineral  matter.  Bermudez  asphalt  melts  from  57  to 
87  deg  cent  and  contains  about  4  per  cent  mineral  matter. 

Asphalts,  Petroleum.  A  "rubberlike"  asphalt  of  almost  any  desired  melting  point  up  to  150  deg 
cent.  The  residue  from  the  distillation  of  asphaltic  or  mixed-base  petroleums.  Sometimes  called 
residual  pitches  or  asphalts.  Have  greater  purity  and  uniformity  than  natural  asphalts.  Used  for 
potting,  impregnating,  etc.  Some  residual  asphalts  weather  very  badly  when  exposed  to  sunlight. 
Petroleum  asphalt  may  be  "blown"  with  air  and  steam  to  oxidize  the  asphalt  partially  and  to  increase 
the  melting  point.  They  also  may  be  modified  by  adding  rosin  derivatives  to  increase  the  fluidity  at 
high  temperatures  without  drop  in  melting  point. 

Asphalts,  Sulfurized.  Sulfur  has  a  hardening  action  on  asphalt,  similar  to  vulcanization  of  rubber 
in  some  respects.  Used  in  corona-resisting  wire  insulation. 

Asphaltites.  Asphaltlike  substances  but  much  harder;  have  melting  points  above  120  deg  cent. 
Not  as  soluble  in  petroleum  hydrocarbons.  The  most  important  varieties  are:  gilsonite,  m.p.  130 
deg  cent;  glance  pitch  or  manjak,  m.p.  160  deg.  cent;  and  graharnite,  m.p.  175  deg  cent  Another 
group  of  asphaltites  containing  so-called  pyrobitumens  are  practically  infusible.  Elaterite,  wurtzilite, 
albertite,  and  impsonite  are  members  of  this  group  and  are  almost  insoluble  in  the  usual  solvents. 
The  first  group  is  used  in  the  manufacture  of  varnishes  and  japans. 

Bakelite.*  Bakelite  Corp.  Phenolic,  cellulose  acetate,  polystyrene,  urea,  and  other  resins,  plastics, 
and  molding  powders. 

Balata.  A  rubberlike  natural  material  similar  to  gutta-percha  and  used  for  similar  purposes.  M.p. 
150  dec  cent.  Dielectric  constant  2.6  to  3.5,  depending  on  purity.  May  be  deresinified  to  improve 
properties  for  use  in  submarine  cables.  See  Kemp,  J.  Franklin  Inst.,  Vol.  211,  No.  1,  p.  37. 

Beeswax.  A  white  to  yellow  insect  wax.  Plastic  at  30  deg  cent.  Melts  from  62  to  64  deg  cent, 
A  good  electrical  insulator.  Has  a  large  negative  coefficient  of  volume  resistivity;  is  bleached  by 
sunlight  and  turns  brown  with  age. 

Beetle.*    American  Cyanamid  Co.    Urea  formaldehyde  molding  powders. 

Buna  N.  Synthetic  rubber  made  from  butadiene  and  acrylonitrile  by  emulsion  polymerization. 
Heat-,  oil-,  and  solvent-resistant,  but  electrical  properties  are  not  good.  Trade  names:  Hycar  OR, 
Perbunan,  Ch&migum. 

Buna  S.  Common  type  of  synthetic  rubber  made  from  butadiene  and  styrene  by  emulsion  polymer- 
ization. Rubber  made  in  government  plants  is  known  as  GR-S.  Insulation  fully  equal  to  that  of 
natural  rubber  in  electrical  properties  can  be  made  from  Buna  S  although  the  mechanical  properties 
are  not  quite  so  good.  Its  general  behavior  is  quite  similar  to  that  of  natural  rubber. 

Butacite.*    E.  I.  du  Pont  de  Nemours  &  Co.     Polyvinyl  butyral  plastics. 

Butvar.*     Shawinigan  Products  Corp.     Polyvinyl  butyral  plastics. 

Butyl.  A  synthetic  rubber  made  from  isobutylene  and  butadiene  by  polymerization  at  low  tem- 
peratures with  a  catalyst.  It  has  excellent  electrical  properties,  good  resistance  to  heat,  ozone,  and 
corona,  and  low  permeability  to  gases  but  is  inferior  to  natural  rubber  in  strength.  When  made  in 
government  plants  it  was  known  as  GR-I. 

Casein.  Prepared  from  skim  milk  by  rennet  treatment  and  hardened  by  soaking  in  formaldehyde. 
Before  hardening  it  can  be  extruded  and  pressed;  after  hardening  it  is  readily  machined.  It  softens  in 
hot  water  at  100  deg  cent  and  can  be  blanked  or  molded  to  a  limited  extent.  It  is  not  a  particularly 
good  electrical  insulator  but  is  occasionally  useful  for  small  parts.  It  is  readily  colored  either  before  or 
after  fabrication. 

Catalin.*     Catalin  Corp.     Cast  phenolic  resins. 

Cellophane.*  E.  I.  du  Pont  de  Nemours  &  Co.  Regenerated  cellulose  film,  lacquered  to  reduce 
moisture  transmission. 

Celluloid.*     Celanese  Plastics  Corp.     Cellulose  nitrate. 

Cellulose  Acetate.  A  thermoplastic  prepared  by  treatment  of  cotton  linters  or  other  cellulose  with 
acetic  anhydride  and  acid  and  the  addition  of  suitable  plasticizers.  Variation  of  processing  and  plasti- 
cizer  gives  a  wide  range  of  molding  flows  and  of  the  heat  resistance,  flow  mobility,  and  other  properties 
of  the  product.  Available  in  all  colors  and  in  the  forms  of  film,  sheet,  rod,  and  molding  powders  for 

*  Trademark  names. 


2-36  PROPERTIES  OF  MATERIALS 

compression  and  injection  molding  or  for  extrusion.  Films  and  sheet  of  electrical  grade  are  non-cor- 
rosive to  fine  copper  wires  even  under  conditions  favorable  to  electrolysis  and  are  superior  for  coil  con- 
struction. Some  molded  materials  contain  volatile  plasticizers  and  shrink  in  time  or  with  heating.  The 
heat  distortion  point  of  many  grades  is  also  quite  low.  Cellulose  acetate  is  not  attacked  by  oils  but  is 
dissolved  by  ketones,  esters,  and  other  solvents.  Specifications  for  sheets:  ASTM  D786;  for  molding 
powders,  ASTM  D706.  Trade  names:  Fibestos,  Lumarith,  Plastacele,  Tenite  I. 

Cellulose  Acetate-butyrate.  Similar  to  cellulose  acetate  except  that  it  is  a  mixed  ester  of  cellulose. 
Less  plasticizer  is  required  to  obtain  molding  flow,  and  heat  resistance  is  improved  slightly.  It  is  widely 
used  in  tape  form  for  wire  insulation  and  as  a  molding  powder.  Specifications  for  molding  powders: 
ASTM  D707.  Trade  name:  Tenite  II. 

Cellulose  Nitrate.  Also  known  as  pyroxylin  and  Celluloid.  It  is  made  by  nitrating  cotton  linters 
or  wood  pulp  and  adding  suitable  plasticizers  such  as  camphor.  Basically  highly  inflammable,  but 
appropriate  plasticizers  reduce  the  inflammability.  Very  tough  and  is  water-  and  chemical-resistant. 
Marketed  as  sheets,  rods,  tubes,  and  molding  compositions.  Specifications  for  sheet,  rod,  and  tube: 
ASTM  D701.  Trade  names:  Celluloid,  Pyralin,  Nitron,  Nixonoid. 

Cellulose  Propionate.  A  new  pkstic  similar  to  cellulose  acetate  but  with  improved  impact  strength 
and  excellent  dimensional  stability.  Good  electrical  properties  and  low  water  absorption.  Trade 
name:  Fcrticel. 

Celeron.*    Continental-Diamond  Fibre  Co.     Macerated-fabric-base  phenolic  moldings. 

Ceramics.  Electrical  ceramics  include  materials  sintered,  fused,  or  fired  at  high  temperatures,  such 
as  porcelain,  steatite,  cordierite,  glass,  glass-bonded  mica,  titanium  and  zircon  ceramics,  q.v.  Ce- 
ramics are  characterized  by  heat  resistance,  permanency  of  dimension,  low  water  absorption,  low  ther- 
mal expansion,  and  excellent  electrical  properties.  Care  is  required  in  mounting  ceramic  parts  since 
the  modulus  of  elasticity  is  high  and  a  slight  bending  develops  high  stresses.  With  the  exception  of 
glass-bonded  mica,  ceramics  are  not  machinable  by  normal  methods  and  are  normally  cast,  molded, 
pressed,  or  extruded  and  machined  in  the  unfired  state  and  then  fired  at  a  high  temperature.  Since 
firing  shrinkage  is  around  12  per  cent,  tolerances  are  not  close,  usually  of  the  order  of  1  per  cent  of  the 
dimension.  A  limited  amount  of  grinding  can  be  done  after  firing. 

Ceresin.  A  white  or  yellow  wax,  with  exceptional  dielectric  properties.  Used  extensively  alone 
and  mixed  with  other  waxes.  M.p.  65  to  67  deg  cent.  '  Soluble  in  oils  and  petroleum  distillates. 
Very  water-resistant  and  has  high  surface  resistivity. 

Cerex.*     Monsanto  Chemicals  Co.     Polystyrene  copolymer. 

Chatterton's  Compound.  Fusible  composition  of  gutta-percha,  rosin,  and  tar  used  in  submarine 
cable  construction  for  sealing  cable  ends,  etc. 

Copaline.*    American  Phenolic  Corp.     Solid-dielectric  r-f  cables. 

Cordierite  Ceramics.  Consist  principally  of  the  crystal  cordierite,  a  magnesium  aluminum  silicate, 
and  are  characterized  by  a  very  low  coefficient  of  thermal  expansion  and  a  power  factor  much  higher 
than  that  of  steatites.  Trade  name:  Alsimag  72  and  208. 

Co-ro-lite.*     Colombian  Rope  Co.     Sisal-fiber-reinforced  phenolic  materials. 

Corprene.*    Armstrong  Cork  Products  Co.     Cork-loaded  neoprene  sheet  in  various  grades. 

Dilectene.*    Continental-Diamond  Fibre  Co.    Aniline  formaldehyde  sheets  and  rods. 

Dilecto.*     Continental-Diamond  Fibre  Co.     Laminated  phenolic  sheet,  rod,  and  tubes. 

Dri-film.*     General  Electric  Co.     Silicone  treatment  for  ceramics. 

Durez.*    Durez  Plastics  and  Chemicals,  Inc.    Phenolic  resins  and  molding  powders. 

Durite.*     Durite  Plastics,  Inc.     Phenolic  and  furfural  resins. 

Ebonite.     Another  name  for  hard  rubber. 

Empire.*  Mica  Insulator  Co.  Varnished  cloth,  sheet,  and  tape.  Silk,  rayon,  Fiberglas,  2  to  40 
mils  thick. 

EnameL  General  term  for  a  substance  producing  a  colored  glossy  coating.  Organic  enamels  are 
made  by  pigmenting  or  coloring  lacquers  or  varnishes,  and  inorganic  enamels  by  pigmenting  low- 
melting  glasses.  (See  discussions  below.) 

Enamel,  Varnish.  Pigmented  varnishes  or  varnishes  colored  by  asphalts  or  colored  resins.  Varnish 
enamels  may  either  dry  in  air  in  4  to  16  hr,  depending  on  the  type,  or  be  compounded  to  dry  by  baking 
from  2  to  8  hr  at  100  to  150  deg  cent.  The  baking  types  usually  have  better  adhesion  and  hardness  and 
higher  dielectric  properties.  The  japans  are  varnish  enamels  made  with  asphalts  for  baking. 

Enamel,  Vitreous.  A  silicate  or  borosilicate  glass  with  the  melting  point  lowered  by  the  addition 
of  various  fluxes,  such  as  metallic  oxides  and  salts.  Some  metallic  oxides  are  used  for  coloring  the  enamel 
and  others  to  increase  the  opacity.  The  enamel  is  usually  applied  in  two  or  more  coats  when  freedom 
from  pmholes  is  desired.  The  same  composition  may  be  used  for  all  coats  in  some  cases,  although 
specially  formulated  ground  coats  are  usually  superior.  The  enamel  coats  are  applied  by  dipping  or 
spraying,  using  a  suspension  of  the  finely  ground  enamel,  known  as  frits,  in  a  clay-and-water  medium. 
For  flat  work  the  powdered  frits  are  sometimes  sieved  onto  the  work  in  the  so-called  dry  process.  The 
coats  are  "fired"  at  temperatures  up  to  850  deg  cent  for  periods  of  3  to  30  min.  The  temperature  and 
time  are  critical  for  a  given  composition  of  enamel  and  for  the  size  of  the  enameled  object.  Some  en- 
amels have  very  satisfactory  dielectric  properties  and  may  even  be  used  for  the  dielectric  of  small  adjust- 
able capacitors,  but  the  major  electrical  use  is  the  manufacture  of  vitreous  enamel  resistors.  For  this 
use  the  formula  is  carefully  adjusted  to  obtain  a  coefficient  of  thermal  expansion  suitable  for  the  wire 
and  coil  forms,  to  prevent  fracture  of  the  wire  with  cycles  of  heating  and  cooling. 

Enamel  Wire.  Enamel  for  insulating  wire  is  usually  a  varnish  enamel.  A  hard,  tough,  and  flexible 
covering  is  produced  with  a  high  dielectric  strength.  The  enamel  thickness  varies  from  0.0001  to 
0.0015  in.,  depending  on  the  wire  size.  Enamel  is  suitable  for  continuous  operation  at  105  deg  cent 
Wire  enamels  vary  slightly  in  resistances  to  oils  and  solvents.  The  better  grades  are  suitable  for  con- 

*  Trademark  names. 


SOLID   DIELECTRIC  MATERIALS  2-37 

tinuous  use  in  mineral  oil  up  to  80  deg  cent;  48  hr  at  105  deg  cent  should  have  no  visible  effect.  Most 
varnish  and  lacquer  solvents  will  attack  wire  enamel  to  some  degree  if  the  contact  is  prolonged.  It  is 
important,  therefore,  to  expel  the  solvents  from  varnish  impregnation  or  cementing  processes  with 
reasonable  promptness.  It  is  necessary  in  most  cementing  processes  for  the  solvents  to  attack  the 
enamel  slightly  in  order  to  secure  a  good  bond.  Wax  impregnation  has  practically  no  effect  on  the 
enamel,  but  impregnation  with  asphaltic  compositions  may  cause  a  severe  attack  on  the  enamel.  The 
enamel  is  moderately  resistant  to  abrasion  and  pressure,  and  wire  may  be  random  wound  for  some  uses 
where  a  few  shorted  turns  are  allowable.  It  is  also  used  successfully  on  core  plates  where  it  is  subjected 
to  considerable  pressure  and  is  occasionally  filled  with  mica  dust  to  improve  the  resistance  to  pressure. 

Ethocel.*     Dow  Chemical  Co.     Ethyl  cellulose  molding  powder. 

Ethofoil.*     Dow  Chemical  Co.    Ethyl  cellulose  film. 

Ethyl  Cellulose.  Prepared  by  replacing  the  hydrogen  of  cellulose  hydroxyl  groups  with  the  ethyl 
group  by  means  of  ethyl  chloride  or  ethyl  sulf  ate.  It  is  thermoplastic  and  is  tough  and  flexible  through 
wide  ranges  of  temperature.  It  is  very  stable  to  heat  and  light  and  has  excellent  electrical  properties. 
It  is  available  in  "hot-melt"  compounds,  in  casting  compositions  (Thermocast  *) ,  in  films,  as  non-rigid 
sheets  and  extrusions,  and  as  molding  compositions.  The  electrical  properties  are  well  retained  under 
moist  conditions  and  actually  may  improve  with  increasing  temperatures,  contrary  to  the  general  rule. 
Specifications  for  non-rigid:  ASTM  D743;  for  molding:  ASTM  D787.  Trade  names:  Ethocel,  Lumartih 
EC,  Chemaco,  Hercules  EC. 

Fiber,  Vulcanized.  Manufactured  by  treating  rag  paper  with  zinc  chloride,  pressing  into  sheets, 
and  thorough  washing.  Fiber  will  absorb  water  up  to  about  60  per  cent  if  immersed  for  a  sufficient 
length  of  time  and  will  increase  to  nearly  double  the  thickness.  Solvents  and  oils  have  practically  no 
effect  on  fiber.  Fiber  has  a  natural  moisture  content  of  5  to  6  per  cent  at  40  to  60  per  cent  relative 
humidity,  which  will  decrease  at  low  humidities  and  increase  at  high  humidities.  Heating  at  80  to 
100  deg  cent  for  long  periods  will  dry  out  and  warp  the  fiber  and  impair  the  flexibility  and  toughness. 
From  100  to  170  deg  cent  the  drying  is  very  rapid,  and  the  fiber  will  become  brittle  in  a  few  hours. 
Above  170  deg  cent  the  material  will  char  on  long  heating.  It  chars  in  a  short  time  at  200  deg  cent.  The 
maximum  safe  operating  limit  is  about  150  deg  cent.  Fiber  is  made  in  gray,  black,  red,  and  white,  but 
there  is  little  difference  in  the  properties  due  to  the  colors  of  the  same  grade  of  stock.  Grade  differences 
are  due  to  selection  of  rags,  increased  pressure  in  order  to  increase  density  and  hardness,  and  more 
careful  processing.  The  highest  grade  is  the  electrical  grade  made  in  thin  sheets  for  use  in  slot  insula- 
tion, etc.,  and  commonly  known  as  "fish  paper."  This  grade  has  exceptionally  high  dielectric  and 
mechanical  strength  and  a  density  of  about  1.3.  Hard  fiber  (bone  fiber,  horn  fiber)  has  a  density  of 
about  1.3,  and  is  considerably  harder  than  the  commercial  grade.  The  commercial  grade  has  slightly 
lower  mechanical  strength  but  about  the  same  dielectric  strength  and  shows  considerably  higher  water 
absorption  on  1-hr  immersion.  A  grade  especially  made  for  forming  and  swaging  contains  glycerin  to 
soften  the  material  by  retaining  moisture.  It  can  be  swaged,  spun,  and  formed  readily  for  bushings, 
grommets,  etc.  All  grades  are  readily  machinable  and  can  be  punched  and  also  formed  dry  to  a  limited 
extent.  For  more  severe  operations,  the  sheets  can  be  soaked  in  water  and  formed  in  heated  dies.  The 
fiber  bends  best  parallel  to  the  grain. 

Fiberglas.*    Owens-Corning  Fiberglas  Corp.     Glass  yarn  and  textiles. 

Fibestos.*     Monsanto  Chemical  Co.     Cellulose  acetate. 

Fibron.*  Irvington  Varnish  and  Insulator  Co.  Flexible  plastic  products.  Polyethylene  tape  and 
tubing. 

Fish  Paper.  Common  name  for  superior  grade  of  thin  vulcanized  fiber,  also  known  as  tarpon  paper, 
leather  paper,  leatheroid,  and  fiberoid.  It  is  usually  supplied  in  a  dark  gray  color. 

Flamenol.*    General  Electric  Co.     Polyvinyl  chloride  wire  insulation. 

Formex.*    General  Electric  Co.     Polyvinyl  formal  magnet  wire  insulation. 

Formica.*    Formica  Insulation  Co.     Phenolic  laminates. 

Formvar.*     Shawinigan  Products  Corp.     Polyvinyl  formal. 

Forticel.*     Celanese  Plastics  Corp.     Cellulose  propionate  plastic. 

Fortisan.*     Celanese  Corp.  of  America.     High-strength  regenerated  cellulose  yarns. 

Fullerboard.    Another  name  for  pressboard. 

Furfural  Resins.  Phenol  and  furfural  react  to  form  resins  similar  to  phenol-formaldehyde  resins 
but  generally  dark  in  color.  They  are  used  for  transfer  molding  and  cold  molding.  They  withstand 
high  temperatures  and  have  comparatively  good  arc  resistance.  Trade  name:  Durite. 

Gelva.*     Shawinigan  Products  Corp.     Polyvinyl  acetate. 

Geon.*    B.  F.  Goodrich  Chemical  Co.     Polyvinyl  chloride  and  vinyl  vinylidene  chloride. 

Gilsonite.     Variety  of  natural  asphalt.     M.p.  122  to  188  deg  cent. 

Glass.  Physically,  glass  is  an  amorphous,  undercooled  liquid  composed  of  silica  and  metallic  sili- 
cates and  hence  has  no  crystal  structure  or  sharp  melting  point.  The  plastic  nature  of  glass  at  elevated 
temperatures  permits  fabrication  by  drawing,  blowing,  and  pressing,  as  well  as  by  casting.  Only  simple 
shapes  can  be  molded.  All  glass  must  be  carefully  annealed  to  prevent  residual  stresses  which  reduce 
the  strength,  or  surface-chilled  in  a  controlled  manner,  as  in  so-caned  tempered  glass,  so  that  the  residual 
stresses  will  increase  the  strength.  Since  glass  has  a  high  elastic  modulus  and  no  internal  structure  to 
interrupt  stress  patterns,  it  is  extremely  sensitive  to  stress  concentrations.  Glass  parts  must  be  care- 
fully designed,  and  surface  damage  must  be  avoided.  Exclusive  of  glasses  designed  for  sealing  purposes, 
lime  glass,  lead  glass,  borosilicate  glass  (Pyrex  types),  and  high-silica  glass  (Vycor)  are  the  principal 
electrical  glasses.  The  borosilicate  and  high-silica  types  have  lower  coefficients  of  thermal  expansion, 
giving  improved  resistance  to  thermal  shock.  Electrical  glasses  have  high  dielectric  strength  and  vol- 
ume resistivity  and  low  power  factor.  These  properties  depreciate  with  rising  temperature,  and  at 
temperatures  from  150  to  200  deg  cent  a  rapid  rise  in  power  factor  and  decrease  in  dielectric  strength 

*  Trademark  names. 


2-38  PROPERTIES  OF  MATERIALS 

begin.  The  high-silica  glasses  are  superior  in  this  respect.  The  surface  of  glass  is  readily  wetted  by 
water  so  that  the  surface  resistivity  is  seriously  reduced  by  relative  humidities  above  70  per  cent.  This 
is  perhaps  the  most  serious  defect  of  glass  as  an  electrical  insulator,  but  recent  work  has  indicated  that 
leakage  may  be  considerably  reduced  by  treatment  with  sih' cones,  q.v.  The  Multiform  process,  in 
which  glass  is  powdered,  pressed  to  shape,  and  fired  like  porcelain,  makes  possible  the  production  of 
complicated  parts  to  tolerances  of  1  to  2  per  cent  and  permits  the  use  of  high-silica  glasses  with  very 
low  losses  and  low  thermal  expansion.  Recently  developed  techniques  permit  the  firing  of  metallic 
coatings  on  glass,  which  may  be  used  as  circuit  conductors,  for  soldered  connections,  or  hermetic  seal- 
ing by  soldering.  By  alteration  of  the  composition  of  glass,  the  thermal  expansion  may  be  matched  to 
certain  metals  so  that  dependable  seals  resistant  to  thermal  shock  can  be  produced.  Lead-through 
seals  are  available  in  single  or  multiple  form,  which  may  be  soldered  in  a  metal  container  to  give  a 
hermetic  seaL 

Glass  Textiles.  Glass  drawn  into  thin  filaments  (2  to  3  X  10~4  in.  diameter)  has  an  enormous  tensile 
strength  (4  to  5  X  105  Ib  per  sq  in.),  owing  to  an  absence  of  shearing  stresses.  Glass  textiles  (Fiber gl 'as  *) 
are  made  from  glass  yarn  of  two  basic  types:  continuous-filament  yarns  made  by  hot  drawing  of  fila- 
ments from  glass  "marbles"  in  a  special  machine,  and  staple  yarns  made  from  staple  fiber  produced  by 
steam  drawing  of  molten  glass  into  fibers  varying  from  4  to  18  in.  in  length.  The  individual  fibers  are 
lubricated  and  combined  into  yarns  of  various  constructions.  Glass  textiles  are  available  as  tapes, 
cloth,  sleeving,  cords,  and  yarns  for  serving  or  braiding.  Outstanding  uses  for  glass  textiles  have  been: 
(1)  high-strength  plastic  laminates;  (2)  fireproof  braiding  for  insulated  wire;  (3)  varnished  glass  cloth; 
(4)  serving  for  magnet  wire.  Advantages  of  glass  textiles  in  these  and  other  services  are  greater  re- 
sistance to  heat,  longer  life,  non-inflammability,  increased  moisture  resistance,  and  high  mechanical 
strength. 

Glass-bonded  Mica.  Ground  mica  bonded  with  a  low-melting  glass,  chiefly  lead  borate,  and  some- 
times with  the  addition  of  cryolite  (sodium  aluminum  fluoride).  The  material  is  hot-pressed  at  600  to 
700  deg  cent  to  the  required  form.  Certain  types  can  be  injection-molded.  Metal  inserts  can  be  molded 
in  place,  in  some  cases  for  hermetic  sealing  purposes.  It  is  also  possible  to  cast  aluminum  around  the 
material  or  to  use  it  as  an  insert  in  plastic  molded  parts.  It  can  be  readily  machined  with  carbide  tools, 
or  with  ordinary  tools  for  small  quantities,  to  close  tolerances.  Glass-bonded  mica  has  excellent  elec- 
trical properties,  good  mechanical  properties,  low  coefficient  of  expansion,  and  stability  in  dimensions 
•up  to  300  deg  cent.  Electrical  properties  do  not  deteriorate  rapidly  with  rising  temperature  or  under 
moist  conditions,  except  that  the  surface  resistivity  of  grades  containing  fluorides  may  be  very  low  at 
high  humidities.  Under  condensation  conditions,  the  fluoride  constituents  are  dissolved  and  may  cor- 
rode metal  parts.  Polishing  and  waxing  the  surfaces  will  eliminate  this  condition  but  will  reduce  the 
arc  resistance.  Special  varieties  of  glass-bonded  mica  (Mycalex  K  *)  are  available  with  controlled  dielec- 
tric constants  between  8  and  20.  Specifications:  Army-Navy  Specification  JAN-I-10,  Grade  L-3  or 
L-4.  Trade  names:  Mycalex,  G.  E.  Mycalex,  Mycroy,  Turx. 

GlyptaL*  General  Electric  Co.  Alkyd  resins  and  products  such  as  varnishes,  cements,  varnished 
cloth,  etc.,  made  therefrom. 

Gummon.*    Garfield  Mfg.  Co.    Asbestos  coal-tar  moldings. 

Gutta-percha.  A  grayish-white  to  brown  plastic  substance  but  not  elastic  like  rubber.  Can  be 
molded  under  pressure  at  60  to  100  deg  cent  and  melts  from  120  to  140  deg  cent.  It  vaporizes  above 
190  deg  cent.  Partly  soluble  in  ether,  carbon  tetrachloride,  benzol,  chloroform,  and  carbon  bisulfide; 
insoluble  in  water.  Can  be  vulcanized  with  sulfur  or  sulfur  chloride  like  rubber,  forming  a  hard  sub- 
stance, but  it  is  nearly  always  used  unvulcanized.  It  is  rather  easily  oxidized  in  the  air  and  becomes 
brittle  and  yellowish  gray.  It  is  principally  used  for  the  insulation  of  submarine  cables  and  ia  generally 
applied  uncompounded  by  a  tubing  machine  or  in  strips  like  rubber.  The  power  factor  of  gutta-percha 
is  mayi'mrim  at  room  temperatures,  so  that  the  dielectric  loss  in  actual  service  at  sea-bottom  tempera- 
tures is  quite  low.  If  the  insulation  is  prepared  with  about  1.5  per  cent  moisture  content,  which  is 
close  to  the  saturation  value  in  sea  water,  the  constants  do  not  change  much  in  service.  The  life  is 
very  satisfactory  under  water  but  is  not  very  satisfactory  in  air. 

Halowax.*    Bakelite  Corp.     Chlorinated  naphthalene  liquids  and  waxes  for  impregnating. 

Hard  Rubber.    See  Rubber,  hard. 

Hemit*    Garfield  Mfg.  Co.     Cold-molded  refractory  materials. 

Herculite.*    Pittsburgh  Plate  Glass  Co.     Tempered  glass  products. 

Hycar.*  Hycar  Chemical  Go.  Synthetic  rubber  in  various  grades,  distinguished  by  suffix  letters 
and  numbers. 

Insurok.*    Richardson  Co.    Phenolic  and  urea  laminates. 
f   Isolantite.*    Isolantite,  Inc.    Steatite  ceramics. 

Jute.  A  long  bast  fiber  employed  in  cordage  and  rough  textiles.  Considerably  used  as  a  filler  and 
core  in  cords  and  cables.  Commercial  jute  often  is  softened  and  rendered  less  brittle  by  impregnation 
with  -mineral  oiL  Jute  loses  its  strength  when  damp. 

Kaolin.  Also  called  china  clay.  An  aluminum  silicate  clay  free  from  iron,  used  in  the  manufacture 
of  white  porcelain.  Valuable  as  a  packing  material  around  heating  coils,  etc. 

Kriston.*    B.  F.  Goodrich  Chemical  Co.    AUyl  monomer. 

Lamicoid.*    Mica  Insulator  Co,    Phenolic  laminates. 

Laminates,  Layers  of  paper,  cloth,  or  glass  cloth  impregnated  with  resin  and  pressed  under  heat 
and  high  pressures.  Resins  are  usually  thermosetting,  but  thermoplastic  resins  have  been  used  Low- 
pressure  laminates  use  resins  which  give  off  little  or  no  gas  or  vapor  during  curing,  and  pressure  just 
sumcient  to  hold  the  mass  in  contact  is  required.  Very  large,  shaped  parts  can  be  produced  by  employ- 
ing inflated  or  evacuated  rubber  bags  to  supply  the  low  pressures  needed.  Knitted  cloths  are  often 
used  to  allow  stretching  where  required.  Contact  laminates  can  be  made  with  still  lower  pressures  (as 

*  Trademark  names. 


SOLID  DIELECTRIC  MATERIALS  2-39 

low  as  1  Ib  per  sq  in.).  Trade  names  of  materials  employed  for  contact  laminating  are:  Laminae,* 
American  Cyanamid  Co.;  Thalids*  Monsanto  Chemical  Co.;  Bakelite  Copolymer  Resins,*  Bakelite 
Corp.;  Selectron,*  Pittsburgh  Plate  Glass  Co.;  Vibrins,*  Naugatuck  Chemical  Co.  The  electrical 
properties  of  many  of  these  low-pressure  laminates  are  excellent.  They  were  employed  for  such  appli- 
cations as  radomes  during  the  war.  Properties  of  sample  laminates:  dielectric  constant  3.54  to  4.59  at 
1  Me;  power  factor  0.0075  to  0.0105  at  1  Me. 

Latex.  An  emulsion  of  rubber,  synthetic  rubber,  or  synthetic  resin  in  water,  depositing  a  solid 
film  on  evaporation. 

Lava.*    American  Lava  Corp.    Mineral  talc  machined  to  shape  and  fired  at  high  temperatures. 

Lavite.*    D.  M.  Stewart  Mfg.  Co.     Steatite  ceramic. 

Lenoxite.*    Lenoxite  Div.,  Lenox,  Inc.     Steatite  ceramic. 

Lignin.  Lignin  is  the  binding  material  in  wood.  Two  types  of  plastic  are  made  from  lignin.  In 
one  type  the  whole  wood  is  used;  steamed  chips  are  exploded  by  sudden  pressure  release  and  are  pressed 
into  boards  under  high  pressure  (Masonite,*  Benalite,*  Masonite  Corp.).  In  the  other  type  the  sepa- 
rated lignin,  usually  a  byproduct  of  paper  manufacture,  is  combined  with  other  materials,  such  as 
amines,  furfural,  or  phenol,  to  form  a  thermosetting  resin  which  maybe  combined  with  various  fillers, 
or  is  used  to  impregnate  paper  which  is  hot-pressed  into  laminated  boards  (Lignolite  *) .  Lignin  also 
is  used  as  an  extender  for  phenol-formaldehyde  molding  compounds. 

Lignolite.*     Marathon  Chemical  Co.     Lignin  plastic  sheets. 

Loalin.*     Catalin  Corp.     Polystyrene. 

Lucite.*  E.  I.  du  Pont  de  Nemours  &  Co.  Methyl  methacrylate  sheet  and  molding  powders,  also 
in  heat-resistant  grades. 

Lumarith.*    Celanese  Celluloid  Corp.     Cellulose  acetate  and  ethyl  cellulose  products. 

Lttstron.*     Monsanto  Chemical  Co.     Polystyrene  molding  powders. 

Magnesium  Oxide.  Compressed  magnesium  oxide  is  used  in  the  insulation  of  heating  units  and, 
in  Europe,  for  heat-resistant  coaxial  conductors  (Pyrotenax  *).  Single  or  multiple  conductor  cables  are 
made  by  packing  magnesium  oxide  preforms  and  the  conductors  inside  a  copper  tube  and  drawing  the 
assembly  to  a  smaller  size.  As  a  r-f  coaxial  line  the  losses  are  higher  than  those  of  polyethylene  insula- 
tion, and  the  ends  must  be  well  sealed  against  moisture.  It  is  electrically  smooth  compared  to  an 
insulator-spaced  air  line. 

Makalot.*     Plastics  Div.,  Interlake  Chemical  Co.     Phenolic  resins  and  compounds. 

Masonite*  Die  Stock.     Masonite  Corp.    Exploded  wood  fiber,  densified  under  high  pressure. 

Melamine-formaldehyde.  Thermosetting  resins  prepared  by  reaction  of  formaldehyde,  melamine, 
and  sometimes  dicyandiamide;  the  latter  two  are  derived  from  calcium  cyanamid.  These  resins  are 
heat-  and  arc-resistant  and  have  excellent  electrical  properties  and  low  water  absorption.  Alpha  cel- 
lulose, chopped  rag,  and  mineral  fillers  are  used  in  various  compounds.  Specifications:  ASTM  D704. 
Resin  trade  names :  M elmac,  Resimene,  Plaskon  Melamine. 

Melamine  Glass  Laminates.  These  laminates  are  characterized  by  high  arc  resistance  and  great 
mechanical  strength.  They  are  also  heat-resistant  and  burn  with  some  difficulty.  The  fumes  from  the 
burning  laminate  are  said  to  be  less  toxic  than  those  from  phenolics ;  hence  these  laminates  were  used 
for  combat-vessel  equipment.  The  high-frequency  properties  are  not  outstanding  and  are  not  con- 
trolled in  production.  Specifications:  Joint  Army-Navy  Spec.  JAN-P-13  Type  GMG. 

Melmac.    American  Cyanamid  Co.     Melanune-fonnalaehyde  resins  and  molding  powders. 

Methacrylates.  These  resins  are  members  of  the  acrylic  or  acrylate  resin  group.  The  most  im- 
portant member  is  methyl  methacrylate.  This  plastic  is  produced  by  the  reaction  of  acetone  and  hy- 
drogen cyanide  to  form  acetone  cyanhydrin,  which  is  allowed  to  react  further  with  methyl  alcohol  to 
produce  methyl  methacrylate  monomer  [CH2  *  C(CE3)COOCHg].  This  monomer  is  polymerized  by 
the  aid  of  peroxide  catalysts  and  heat.  The  polymers  are  characterized  by  great  optical  clarity,  light 
transmission  of  92  per  cent,  high  refractive  index  of  1.48  to  1.51,  stability  to  light  and  weather,  and 
good  mechanical  and  electrical  properties.  Arc  resistance  is  high;  vapor  from  the  plastic  actually  tends 
to  quench  arcs.  Power  factor  and  dielectric  constant  decrease  with  increasing  temperature  and  fre- 
quency instead  of  exhibiting  the  normal  increase.  Methyl  methacrylate  in  common  with  other  thermo- 
plastics has  a  low  heat  distortion  point.  Heat-resistant  grades  are  available  that  will  withstand  boil- 
ing in  water.  Specifications  for  sheet,  rods,  and  tubes:  ASTM  D702;  for  molding  compounds:  ASTM 
D788.  Trade  names:  Plexiglas,  Lucite. 

Mica.  A  group  of  natural  complex  aluminum  silicates  with  highly  developed  basal  cleavage  into 
thin,  tough,  flexible  laminae.  It  is  probable  that  if  sufficient  care  were  taken  it  could  be  split  into 
thickness  approaching  molecular  dimensions.  Owing  to  the  fact  that  blocks  are  very  expensive,  mica 
is  usually  split  and  punched  into  parts  for  capacitors  and  spacers.  For  other  uses  the  flakes  are  cemented 
together  with  adhesives  to  make  "built-up"  or  "pasted"  mica,  which  forms  flexible  or  rigid  sheets,  de- 
pending on  the  binder.  Flake  or  dust  mica  is  combined  with  resins  or  glasses  to  form  simple  molded 
shapes  as  well  as  rods  and  sheets.  There  are  several  varieties  of  mica,  including: 

Biotite — iron  mica,  black  mica  Muscovite — potassium     mica, 

Paragonite — sodium  mica  common  mica 

Lepidolite — lithium  mica  Phlogopite — magnesium  mica, 

Lepidomelane — iron  mica  rhombic  mica 

but  only  the  last  two  are  used  for  electrical  work.  Muscovite  comes  in  three  colors:  white  and  ruby, 
both  of  which  are  superior  grades,  and  green,  which  is  inferior  electrically  and  mechanically  to  clear 
grades  of  white  and  ruby.  Muscovite  mica  is  in  general  superior  electrically  and  mechanically  to 
phlogopite,  but  phlogopite  has  superior  heat  resistance.  Phlogopite  ranges  from  a  deep  amber  color  to 
dark  amber  and  milky  white.  It  is  not  so  readily  split  as  muscovite;  it  is  softer,  and  is  lower  in  mechani- 

*  Trademark  names. 


2-40  PROPEBTIES  OF  MATERIALS 

cal  strength.  Phlogopite  mica  does  not  lose  water  up  to  temperatures  of  800  to  900  deg  cent,  and  some 
grades  will  resist  1200  deg  cent  without  complete  disintegration.  For  this  reason  it  is  valuable  for 
spacing  the  elements  in  vacuum  tubes  and  in  heating  devices.  The  maximum  operating  temperature 
is  best  limited  to  1000  deg  cent,  and  that  of  muscovite  to  500  deg  cent.  The  power  factor  and  resistivity 
of  phlogopite  are  much  worse  than  those  of  muscovite,  although  the  dielectric  strength  is  nearly  the 
same.  Stained  muscovite  and  all  phlogopite  are  unsuitable  for  use  in  capacitors  where  low  power  factor 
is  required.  Mica,  in  general,  does  not  decrease  in  dielectric  strength  with  frequency  as  fast  as  most 
dielectrics.  This  fact,  together  with  its  low  power  loss,  enables  carefully  designed  and  built  mica  capac- 
itors to  operate  at  extremely  high  frequencies.  The  defects  occurring  in  mica  are  air  bubbles,  _  stains, 
and  spots.  Mica  is  graded  according  to  freedom  from  defects  as  follows:  highest-grade^  mica  is  clear 
and  free  from  all  defects;  second  highest  grade  has  air  bubbles  between  laminae;  stained  mica  sometimes 
has  some  iron  stains  present;  spotted  mica  is  badly  stained  and  usually  has  inclusions  of  other  minerals. 
Mica,  splittings  are  graded  for  size  according  to  the  largest  usable  rectangular  area.  For  grad- 
ing methods,  see  ASTM  D351.  The  dielectric  strength  of  mica  is  considerably  reduced  by  air  or  mois- 
ture between  the  laminae,  but  the  flexibility  is  somewhat  increased.  Specifications  for  block  mica  and 
films:  ASTM  D748;  for  electrical  tests:  ASTM  D351. 

Mica,  Reconstructed  or  Pasted.  Flake  mica  is  bound  and  pressed  together  with  shellac,  gum, 
asphalt,  or  synthetic  resin  varnishes  and  milled  to  thickness  to  form  sheets  which  may  be  punched  and 
sheared.  Some  grades  are  flexible  and  may  be  formed  to  a  limited  extent  cold.  Others  use  a  thermo- 
plastic binder  and  can  be  formed  to  quite  intricate  shapes  at  100  deg  cent.  Hard  grades  contain  as 
little  as  3  per  cent  binder  and  do  not  compress  appreciably.  See  also  Mica  cloth  and  Mica  paper. 
Specifications  for  materials:  NEMA  Standards  39-55;  for  testing  methods:  ASTM  D352. 

Mica  Cloth.  A  composite  insulation  of  high  dielectric  strength  used  for  insulating  transformers  and 
field  windings. 

Mica  Paper.  Flake  mica  cemented  between  sheets  of  glassine,  rice,  kraft,  or  express  paper.  Mica 
also  is  combined  with  asbestos  paper  or  fibers  to  form  composite  insulations. 

Mica  Plate.     Another  name  for  reconstructed  mica  sheets. 

Micabond.*     Continental-Diamond  Fibre  Co.     Reconstructed  mica  tape,  tubes,  and  sheets. 

Micanite.*     Mica  Insulator  Co.     Reconstructed  mica  products. 

Micarta.*    Westinghouse  Electric  Corp.    Phenolic  laminates. 

Minerallac.*    Minerallae  Electric  Co.     Fusible  asphalt  compounds. 

Molded  Compounds.  Hot-molded  products  are  formed  in.  molds  or  platens  heated  to  a  temperature 
sufficient  to  cause  the  binder  to  flow,  cementing  the  particles  and  producing  a  pure  smooth  binder 
surface  which  lowers  water  absorption  and  increases  surface  resistivity.  Compounds  in  which  binder 
hardens  under  heat  are  called  therm osetting;  those  in  which  binder  becomes  plastic  are  called  thermo- 
plastic, and  molds  must  be  cooled  before  the  article  is  removed.  Some  thermosetting  binders  are 
synthetic  resins  of  the  phenol  formaldehyde,  urea  formaldehyde,  or  melamine  type.  Some  thermo- 
plastic binders  are  shellac,  cellulose  nitrate  and  acetate,  vinyl  resins,  mixtures  of  asphalts  and  hardened 
rosinr  copals,  casein  resins,  sulfur  chloride  phenol  resins,  cumarin  resins,  polystyrene,  and  methacrylate. 
Fillers  may  be  either  fibrous  or  powdered,  wood  flour  being  the  commonest.  Cotton,  silk  flock,  or 
threads  are  used  to  improve  resistance  to  impact,  and  macerated  cloth  to  give  high  impact  resistance, 
with  thennosetting  compounds.  Asbestos  is  used  for  heat-resistant  products,  mica  to  obtain  low 
power  factor,  and  ground  flint,  china  clay,  silex,  stone,  etc.,  to  cheapen  the  article.  Compounds  can 
be  hot-molded  in  great  varieties  of  shapes  with  thin  sections,  metallic  inserts,  threads,  etc.  Tolerances 
can  be  held  to  within  plus  or  minus  0.003  in.  per  in.  when  not  depending  on  mold  closure.  Cold-molded 
products  are  formed  under  pressure  and  subsequently  baked  for  periods  of  from  a  few  hours  to  a  week 
at  temperatures  of  150  to  300  deg  cent.  There  is  little  flow  of  the  binder,  and  the  surface  is  not  very 
smooth  and  depends  on  the  fineness  of  the  filler.  Pieces  distort  slightly  in  baking,  and  the  accuracy 
of  dimensions  is  much  lower  than  in  the  hot-molded  process:  it  amounts  to  plus  or  minus  0.009  to 
0.015.  Binders  are  thick  varnishes,  asphalts,  tung  and  linseed  oils,  anthracene  oils,  and  various  gums 
or  varnish  resins  in  suitable  solvents.  Cold-molded  compounds  are  subdivided  into  refractory  and 
non-refractory,  according  to  the  degree  of  heat  resistance.  In  general,  close  tolerance,  loose  mold 
pieces,  and  threaded  parts  increase  molding  cost.  For  low  cost  the  parts  should  be  designed  with 
adequate  radii,  and  with  no  projections,  indentations,  or  holes  which  require  loose  pieces  in  the  molds. 

Multiform  Glass.*     Corning  Glass  Works.    Powdered  glass,  pressed  and  fired. 

Muscovite.     Variety  of  mica  suitable  for  electrical  insulation.     See  Mica. 

Mycalex.*     Mycalex  Corp.     Glass-bonded  mica,  sheet,  rod,  and  molded. 

Mycalex,*  G.  E.     General  Electric  Co.     Glass-bonded  mica,  sheet,  rod,  and  molded. 

Mycroy.*    Electronic  Mechanics,  Inc.     Glass-bonded  mica. 

Neoprene.  Produced  by  emulsion  polymerization  of  chloroprene,  which  is  derived  from  acetylene. 
Polychloroprene  is  the  chemical  name  for  neoprene  but  neoprene  generally  is  used  even  though  it  is" 
the  trade  name  for  the  E.  I.  du  Pont  de  Nemours  &  Co.  product.  Neoprene  can  be  vulcanized  like 
rubber  but  sulfur  is  not  always  required.  There  are  many  types  of  neoprene  distinguished  by  suffix 
letters ;  some  of  the  types  are  copolymers  with  nitriles.  Neoprene  made  in  government  plants  was  called 
GR-M;  freeze-resistant  neoprene,  FR;  general  purpose,  GN;  low  oil  swelling  and  low  gas  diffusion,  ILS, 
etc.  All  types  are  outstanding  in  oil,  ozone,  and  sunlight  resistance  and  will  not  support  combustion 
Neoprene  may  be  compounded  to  give  power  factors  of  1  per  cent  and  resistivities  of  10 12  ohm-cm  but 
many  compounds  are  poor  electrically.  Neoprene  may  also  be  formulated  for  high  heat  resistance  or 
for  good  abrasion  resistance.  One  application  of  great  value  is  the  jacket  of  portable  cords  and  cables 
Neoprene  also  may  be  made  with  low  specific  resistivity  for  use  as  electrostatic  shields  or  for  potential 
control.  It  also  is  an  excellent  gasket  material.  puwuuai 

Nitrocellulose.     See  Cellulose  nitrate. 

*  Trademark  names. 


SOLID  DIELECTRIC  MATERIALS  2-41 

Nitron.*     Monsanto  Chemical  Co.     Cellulose  nitrate. 

Nixonite.*     Nixon  Nitration  Works.     Cellulose  acetate. 

Nixonoid.*     Nixon  Nitration  Works.     Cellulose  nitrate. 

Nylon.*    E.  I.  du  Pont  de  Nemours  &  Co.     Polyamide  products  of  all  kinds. 

Ozokerite.  A  natural  mineral  wax,  amorphous  and  black  to  dark  brown.  When  bleached,  it  is 
white  to  yellow  or  brown.  Melting  point  is  70  to  80  deg  cent.  The  purified  wax  is  known  as  ceresin. 

Panelyte.*     St.  Regis  Paper  Co.     Phenolic  laminates. 

Paper.  Paper  for  insulating  purposes  should  be  as  free  as  possible  from  all  chemicals  and  from 
conducting  particles  and  should  be  strong  mechanically.  Paper  changes  its  moisture  content  very 
rapidly  with  changes  of  atmospheric  humidity.  Single  sheets  of  thin  paper  will  come  to  equilibrium 
in  as  little  as  15  minutes,  so  that  all  testing,  both  mechanical  and  electrical,  is  done  best  under  condi- 
tions of  controlled  humidity  and  temperature.  The  principal  types  of  paper  of  interest  are  those  fol- 
lowing. Rag  papers  made  with  a  minimum  of  chemicals  and  short  "cooks"  give  strong  paper  with 
heat  resistance  somewhat  improved  over  that  of  chemical  wood  papers.  Manila  papers,  made  from 
manila  fiber  or  from  old  rope,  etc.,  sometimes  with  cotton  or  linen  rags,  have  been  the  standard  cable 
papers  for  some  years  because  of  high  mechanical  and  electrical  properties.  Kraft  papers  can  now  be 
made,  however,  with  equal  or  superior  properties ;  kraft  papers  properly  made  are  very  strong,  have 
excellent  dielectric  properties,  and  are  cheaper  than  rag  or  manila  paper.  Glassine  or  onion-skin  paper 
is  a  highly  beaten  sulfite  stock  with  fair  mechanical  and  electrical  properties.  It  does  not  take  impreg- 
nation well,  however.  High-density,  well-beaten,  heavily  calendered  stocks  in  the  thicker  sizes  are 
much  used  as  insulating  strips.  See  Pressboard.  Material  intermediate  between  paper  and  boards  in 
density  or  thickness  is  known  by  various  names — express  paper  (chemical  wood  fiber),  rope  paper 
(from  old  ropes),  etc. 

Paragutta.  Submarine  cable  insulation  compounded  of  purified  gutta  hydrocarbon  and  deresinified 
rubber. 

Perbunan.*  Standard  Oil  Co.  of  N.  J.  Synthetic  rubber:  a  copolymer  of  butadiene  and  acrylo- 
nitrile. 

Petrolatum.  Comes  in  liquid,  soft,  and  hard  grades.  The  soft  form  is  similar  to  vaseline  and  is 
used  extensively  as  a  paper-cable-impregnating  material.  M.p.  50-55  deg  cent.  The  electrical  prop- 
erties vary  with  purity.  ^J 

Phenol  Fiber.     General  term  for  paper-base  phenolic  laminates. 

Phenolic  Insulating  Materials.  Obtainable  in  two  principal  forms:  molded  parts;  and  laminated 
sheets,  rods,  or  tubes  with  paper  or  fabric  base.  For  molding  compounds,  the  resins  are  combined 
with  the  desired  fillers,  either  by  working  on  rolls  or  by  coating  the  filler  particles  with  a  varnish  and 
drying  (see  discussion  of  molded  compounds).  For  laminated  products,  sheets  of  paper  or  fabric  are 
coated  or  impregnated  with  varnish  and  pressed  hydraulically  between  heated  platens.  Rods  are 
wound  on  small  mandrels  which  are  removed  and  the  roll  is  cured  in  a  mold.  This  leaves  a  weak  center 
section,  and  for  some  purposes  rod  turned  from  sheet  stock  is  preferred,  although  it  does  not  machine 
as  well  and  splits  more  readily.  For  molded  tubes  the  impregnated  paper  is  wound  on  mandrels  and 
the  assembly  cured  in  heated  molds.  Since  the  pressure,  of  course,  is  not  radial  unless  an  expanding 
mandrel  is  used,  the  seams  in  molded  tubes  are  weak  and  tend  to  split  apart.  To  overcome  this,  a 
rolled  tubing  is  manufactured  which  is  cured  by  heated  rolls  during  the  winding.  Since  the  pressure  is 
limited  in  this  process,  the  electrical  properties  generally  are  not  equal  to  molded  tubing  but  the  me- 
chanical properties  are  superior.  Laminated  phenolic  insulation  is  hard,  tough,  and  rigid,  but  more 
elastic  than  equivalent  molded  compounds.  It  is  infusible  and  resists  temperatures  up  to  125  deg  cent, 
but  becomes  slightly  more  brittle  upon  cooling  after  continuous  operations  above  90  deg  cent,  and 
usually  shrinks  somewhat  more  than  it  had  expanded.  A  slight  softening  is  noted  while  the  compound 
is  hot,  of  which  advantage  is  taken  to  reduce  breakage  in  punching  operations.  Stress  applied  while 
hot  causes  a  slight  permanent  set,  and  a  limited  forming  is  thus  possible.  The  dielectric  properties 
are  not  so  good  as  those  of  hard  rubber,  but,  mechanically,  phenolic  insulating  compounds  are  superior 
and  do  not  corrode  metals  or  deteriorate  with  age.  Specifications  for  laminates:  NEMA  Standards, 
and  ASTM  D709;  for  molding  compounds  ASTM  D700. 

Phenolic  Resins.  Phenol  and  various  other  phenolic  substances,  such  as  cresol,  will  condense  and 
polymerize  with  aldehydes  under  the  influence  of  heat  and  a  suitable  catalyst.  Formaldehyde  and 
hexamethylene  tetramine  are  the  commonest  substances  employed  to  react  with  phenols  or  cresols. 
The  reaction  is  catalyzed  by  ammonia,  alkalies,  acids,  and  other  agents.  The  reaction  proceeds  in  two 
or  more  stages.  In  the  first  stage  the  resin  is  fusible  and  soluble  in  acetone  and  other  solvents.  Upon 
further  heating  the  resin  becomes  infusible  and  practically  insoluble.  This  second  stage  is  the  base  of 
thermosetting  molding  compounds  and  of  some  phenolic  laminating  or  baking  varnishes. 

Phenolite.*     National  Vulcanized  Fibre  Co.    Phenolic  laminates. 

Phlogopite.     Variety  of  mica,  q.v. 

Piccolastic.*     Pennsylvania  Industrial  Chemical  Corp.     Substituted  styrene  polymers. 

Plaskon.*     Plaskon  Div.,  Libby-Owens-Ford  Glass  Co.     Urea  or  melamine  molding  compounds. 

Plastacele.*    E.  I.  du  Pont  de  Nemours  <fc  Co.     Cellulose  acetate  products. 

Plax.*    Plax  Corp.    Polystyrene. 

Plexiglas.*    Rohm  and  Haas  Co.     Methyl  methacrylate  products. 

Pliolite.*    Goodyear  Tire  and  Rubber  Co.     Cyclicized  rubber  thermoplastic  resin. 

Polectron.*     General  Aniline  &  Film  Corp.     Polyvinyl  carbazole  resin. 

Polyamides.  Thermoplastic  resins  formed  from  dibasic  acids  and  diamines.  Nylon,  the  most 
important,  is  formed  from  adipic  acid  and  hexamethylene  diamine.  It  is  characterized  by  extraordinary 
strength  and  toughness,  and  by  a  high  degree  of  resistance  to  solvents  and  chemicals.  The  electrical 
characteristics  of  nylon  are  good  but  not  so  outstanding  as  its  mechanical  properties.  It  has  been  used 

*  Trademark  names. 


2-42  PKOPBRTIES  OF  MATERIALS 

successfully  for  thin-wall  coil  forms  and  for  thin-wall  jacketing  of  assault  wire.  Its  strength  and  fungus- 
resisting  properties  have  led  to  its  use  for  many  military  applications  in  connection  with  parachutes, 
aircraft,  guy,  mooring,  and  tow  ropes,  cords,  etc.  It  is  available  as  yarn,  monofilament,  and  molding 
compound.  Yarn  and  monofilament  are  cold-drawn  or  orientated  to  effect  a  very  considerable  increase 
in  tensile  strength.  Specifications:  ASTM  D789. 

Polydichlorostyrene.  Thermoplastic  prepared  by  polymerization  of  dichlorostyrene.  It  is  similar  in 
most  respects  to  polystyrene  except  that  a  considerable  increase  in  heat  distortion  temperature  has 
been  made  with  only  a  slight  sacrifice  in  electrical  properties.  Trade  name:  Styramic  H.T. 

Polyethylene.  Prepared  by  the  polymerization  of  ethylene,'  polyethylene  is  outstanding  for  low 
electrical  losses  at  high  frequencies  and  has  found  extensive  application  as  a  dielectric  in  r-f  coaxial 
cables.  In  thin  sections  it  is  flexible,  but  thick  sections  are  rigid  and  can  be  machined.  It  is  thermo- 
plastic and  is  fabricated  by  injection  or  extrusion.  It  is  available  in  the  forms  of  molding  compound, 
tape,  tubing,  monofilament,  and  rods  or  slabs.  English  practice  includes  plasticizing  with  polyiso- 
butylene,  which  lowers  the  cold-brittleness  point.  Polyethylene  is  insoluble  in  common  solvents  when 
cold  but  dissolves  in  hot  hydrocarbons.  Coatings  of  polyethylene  may  be  applied  by  flame-spraying 
or  by  the  use  of  emulsions.  Coatings  have  a  low  moisture  permeability.  Trade  name:  Polythene. 

Polyflex.*     Plax  Corp.     Flexible  polystyrene  sheet. 

Polystyrene.  Thermoplastic  resin  produced  by  polymerization  of  monomeric  styrene  with  heat  and 
sometimes  a  catalyst  such  as  a  peroxide.  Polystyrene  is  outstanding  for  low  dielectric  loss  at  high 
frequencies.  It  has  high  dielectric  strength  and  good  arc  resistance.  It  has  zero  water  absorption  and 
good  mechanical  strength,  and  it  does  not  become  more  brittle  at  low  temperatures.  It  has  exceptional 
optical  clarity  and  high  refractive  index.  Unfortunately,  it  cannot  be  used  at  temperatures  much  above 
65  deg  cent  without  cold  flow  occurring,  and  some  tendency  for  surface  crazing  exists.  Crazing  may  be 
minimized  by  suitable  heat  treatment  to  remove  surface  strains.  Attempts  to  increase  the  operating 
temperature  by  the  use  of  fillers  have  not  been  too  successful,  for  they  increase  the  tendency  to  crack. 
The  usual  method  of  fabrication  is  injection  molding,  although  many  parts  are  machined  from  sheet, 
rod,  or  tube  stock.  Polystyrene  may  be  drawn  or  oriented  to  form  a  flexible  sheet  (Potyflex  *)  or 
plasticized  to  make  films  for  use  in  capacitors,  etc.  Specifications :  ASTM  D703.  Trade  names :  Loalin, 
Lustron,  Plax,  Pdyflex,  Styron* 

Polystyrene,  Modified.  Polystyrene  has  been  combined  with  chlorinated  diphenyl  to  form  a  non- 
inflammable  plastic  with  a  heat  distortion  point  somewhat  higher  than  that  of  polystyrene.  It  is  also 
easier  to  machine,  but  the  electrical  losses  are  slightly  higher.  Trade  name:  Styramic.  Other  modi- 
fications are  possible,  such  as  copolymerizing  styrene  with  other  materials  such  as  butadiene.  With 
about  25  per  cent  styrene  Buna  S  rubber  is  formed,  but  if  the  styrene  is  considerably  in  excess  a  semi- 
flexible  thermoplastic  (Styraloy  *)  is  produced.  This  material  has  good  dielectric  properties  and  is 
very  tough.  It  is  suitable  for  wire  and  cable  insulating.  Cerex  *  is  another  recently  introduced  styrene 
copolymer  with  improved  heat  resistance,  high  strength  and  hardness,  and  unusual  chemical  resistance, 
but  somewhat  higher  losses  than  polystyrene. 

Polytetrafluoroethylene.  Manufactured  by  polymerizing  gaseous  tetrafluoroethylene;  a  plastic 
with  remarkable  heat,  chemical,  and  solvent  resistance.  Polytetrafluoroethylene  is  very  tough  over  a 
wide  temperature  range  and  has  a  loss  factor  less  than  that  of  polystyrene;  its  dielectric  constant  of 
2.0  is  the  lowest  of  any  solid  insulating  material.  It  is  very  expensive  as  of  1949,  and  extrusion  or  mold- 
ing is  slow.  Some  machining  is  necessary  on  most  parts  since  molding  is  very  difficult.  Trade  name: 
Teflon^ 

Polythene.*    E.  I.  du  Pont  de  Nemours  &  Co.    Polyethylene. 

Polyvinyl.     See  Vinyl. 

Porcelain.  A  ceramic  body  usually  composed  of  clay,  feldspar,  and  flint,  finely  ground,  mixed 
with  water,  formed  to  desired  shape,  dried,  and  fired  at  temperatures  usually  ranging  from  1300  to 
1800  deg  cent.  When  desired,  glaze  is  applied  on  all  surfaces,  except  the  base  on  which  the  part  rests 
in  the  kiln  during  firing,  by  painting  on  a  composition  which  fuses  to  a  translucent  or  transparent  glass. 
Mixtures  of  china  clay  or  kaolin,  which  are  slightly  plastic,  and  ball  clay,  which  is  very  plastic,  are 
used  to  give  the  necessary  working  properties  to  the  wet  dough.  Feldspar  is  a  naturally  occurring 
potassium  aluminum  silicate.  Flint  is  added  in  the  form  of  ground  sand  or  quartz.  Normal  por- 
celains contain  from  20  to  60  per  cent  of  clay,  from  15  to  50  per  cent  of  feldspar,  and  from  0  to  65  per 
cent  of  flint.  Magnesia  sometimes  is  added  up  to  50  per  cent  to  improve  the  strength  at  high  tempera- 
tures. Special  porcelains  vary  widely  in  composition.  Some  are  made  from  natural  aluminum  silicates 
such  as  andalusite  with  enough  clay  to  give  a  bond.  Magnesium  silicate  ceramics  made  with  bases  of 
talc,  steatite,  etc.,  give  modified  porcelains  with  superior  electrical  and  mechanical  properties.  The 
raw  "body"  or  dough  is  formed  into  shape  by  two  distinct  processes.  In  the  dry  process  the  mass  is 
compressed  in  steel  dies.  Parts  must  have  "draft"  and  taper  similar  to  die  castings,  and  a  tolerance  of 
plus  or  minus  1/64  in.  per  in.  is  necessary  to  allow  for  shrinkage  variation  in  firing  and  wear  of  molds  by 
abrasion.  Minimum  commercial  tolerance  on  thickness  is  plus  or  minus  0.010  in.  Dry-process  por- 
celain is  used  for  insulation  under  5000  volts  only,  since  it  is  porous  and  weaker  mechanically  and 
electrically  than  wet-process  porcelain.  The  porosity  is  from  3  to  5  times  as  high  as  that  of  wet-process 
parts.  Wet-process  porcelain  is  made  by  forming  the  dough  to  the  approximate  shape,  drying  and 
machining  on  vertical  or  horizontal  lathes  to  final  shape.  It  also  is  sometimes  cast  in  a  fluid  state  in 
absorbent  molds  which  remove  enough  water  to  enable  the  part  to  be  removed  after  some  time  and 
dried.  Cast  porcelain  compares  favorably  with  formed  wet-process  porcelain.  Wet-process  porcelain 
has  a  very  low  porosity  and  high  dielectric  strength.  A  tolerance  of  about  l/32  in.  per  in. ,  plus  or  minus 
is  necessary  for  commercial  manufacture.  All  porcelain  is  relatively  weak  in  tensile,  flexural  and 
impact  strength.  It  has  poor  resistance  to  thermal  shock  except  in  special  grades.  The  mechanical 
strength  depends  upon  the  flint  content,  the  heat  resistance  upon  the  clay,  and  the  dielectric  strength 

*  Trademark  names. 


SOLID  DIELECTRIC  MATERIALS  2-43 

upon  the  feldspar,  which  unfortunately  tends  to  make  the  parts  brittle.  Porcelain  is  comparatively 
inexpensive  and  chemically  inert.  Dry-process  parts  are  considerably  cheaper  than  wet-process  parts. 
Glazing  improves  the  resistance  to  weathering,  but  it  must  have  a  coefficient  of  expansion  similar  to 
that  of  the  body,  for  a  cracked  or  "crazed"  glaze  reduces  the  strength.  The  insulation  resistance  of 
normal  porcelains  drops  rapidly  above  300  deg  cent;  to  minimize  this  effect,  alkali  metals  are  reduced 
in  amount  as  much  as  possible.  For  high-temperature  work  free  quartz  in  the  fired  body  is  undesirable 
since  it  has  irregularities  in  its  thermal  expansion  curve  which  tend  to  cause  cracking,  and  so  it  is 
eliminated  as  far  as  possible.  The  desirable  structure  is  usually  crystals  of  aluminum  silicates  (known 
as  mullite  and  sillimanite)  evenly  embedded  throughout  a  glassy  matrix.  Pores  are,  of  course,  highly 
undesirable.  Fired  porcelain  parts  can  be  ground  to  meet  <slose  tolerances,  and  two  or  more  subparts 
can  be  fastened  together  with  neat  Portland  cement,  litharge-glycerin  cement,  or  asphalt  and  resin 
base  compounds.  Low-melting  metals,  such  as  babbitt,  may  be  cast  around  porcelain  with  some 
attendant  danger  of  cracking. 

Pressboard.  A  material  similar  to  paper  except  that  it  is  thicker,  less  flexible,  and  usually  denser. 
Grades  containing  above  50  per  cent  cotton  fiber  may  be  formed  by  heat  and  pressure  into  simple 
shapes.  The  better  grades"  are  also  known  as  fullerboard.  Pressboard  is  much  used  for  low-frequency 
coil  construction  with  subsequent  impregnation.  The  material,  of  course,  is  hygroscopic  and  must  be 
treated  with  oil,  wax,  varnish,  or  other  compounds  to  increase  dielectric  strength  and  repel  moisture. 
The  impregnated  material  is  a  cheap  and  satisfactory  insulator  where  the  highest  dielectric  properties 
are  not  required. 

Prestite.*    Westinghouse  Electric  Corp.     Special  dry-process  porcelain. 

Pyralin.*    E.  I.  du  Pont  de  Nemours  &  Co.    Cellulose  nitrate. 

Pyrex.*    Corning  Glass  Works.     Electrical,  heat,  and  chemical  resistant  glasses. 

Pyroxylin.     See  Cellulose  nitrate. 

"Q"  Max**    Communication  Products  Co.    Low-loss  r-f  coil  lacquer, 

Quartz,  Fused.  Silicon  dioxide  fused  at  1750  deg  cent  to  a  clear,  translucent,  glassy  mass.  Very 
stable,  will  not  absorb  water,  and  is  an  exceptional  insulator.  Strong  mechanically  and  extremely 
resistant  to  thermal  shock  on  account  of  the  low  coefficient  of  expansion*  It  is  not  attacked  by  solvents 
or  solutions  except  by  hydrofluoric  acid  and  slowly  by  concentrated  alkalies.  For  high  temperatures  it 
must  be  kept  clean,  as  traces  of  metallic  salts  or  oxides  will  flux  the  quartz  to  form  a  low-melting  glass 
and  cause  failure  of  the  tube  or  other  device.  It  is  available  in  rods,  blocks,  tubes,  and  extruded  shapes. 
Special  shapes  can  be  cast  in  graphite  molds.  It  can  be  ground  and  disk-sawed  readily  since  it  does  not 
•crack  easily.  It  is  very  expensive. 

Rayon.  The  three  common  types  of  rayon  are  acetate,  viscose,  and  cuprammonium.  Viscose 
rayon  is  the  strongest  and  most  suitable  for  protective  braids  on  hook-up  wire  but  is  not  so  resistant  to 
abrasion  as  cotton.  Viscose  rayon  contains  traces  of  residual  sulfur  which  may  cause  corrosion  if  it  is 
used  for  magnet  wire  insulation;  acetate  and  cuprammonium  rayon  are  suitable  for  such  applications. 
Acetate  rayon  is  especially  suitable  for  use  with  very  fine  wires  for  it  is  non-corrosive  even  under  condi- 
tions where  electrolysis  "would  take  place. 

Resimene.*    Monsanto  Chemical  Co.     Melamine  molding  compounds. 

Resinox.*     Monsanto  Chemical  Co.    Phenolic  molding  compounds. 

Resistoflex.*    Resistoflex  Corp.     Polyvinyl  alcohol  products. 

Rosin.  Rosin  is  a  natural  resin  obtained  by  steam  distillation  of  turpentine  and  rosin  oils  from 
the  exudations  of  certain  varieties  of  pine  trees.  Rosin  comes  in  letter  grades.  WW  (water  white)  is 
the  best,  descending  in  reverse  alphabetical  order  to  A  and  B,  which  are  very  impure  grades,  contain- 
ing much  dirt,  and  almost  black.  It  is  universally  graded  by  the  color.  Although  the  properties  do  not 
vary  directly  with  the  color,  the  color  is  important  for  use  in  varnishes.  Rosin  in  grades  from  WW  to 
H  is  extensively  used  in  oil  and  wax  impregnating  compounds.  It  also  is  very  cheap.  Probably  the 
most  important  use  in  the  electrical  industry  is  non-corrosive  soldering  flux.  Solutions  of  rosin  in 
No.  1  S.D.  alcohol,  or  in  alcohol  and  ethyl  acetate,  form  a  substantially  non-corrosive  soldering  flux 
and  yet  the  activity  of  rosin  at  the  temperature  of  melted  solder  is  sufficient  to  remove  thin  coats  of 
metallic  oxides  and  insure  a  good  joint  on  tin,  copper,  brass,  or  nickel  silver.  It  is  unsatisfactory  for 
steel.  Any  rosin  left  around  the  joint  is  non-conductive,  which  is  a  further  advantage. 

Rubber,  Cyclicized,  Thermoplastic  resin  derived  from  natural  or  special  synthetic  rubber  by  treat- 
ment with  stannic  chloride  or  chlorostannic  acid.  This  resin  is  extremely  resistant  to  moisture  diffu- 
sion and  may  be  added  to  wax  mixtures  to  decrease  cracking  at  low  temperatures.  Trade  names: 
Pliolite,  Marlon  B. 

Rubber,  Hard.  Hard  rubber  is  usually  vulcanized  with  20  to  30  per  cent  sulfur,  in  the  form  of 
sheets,  rodsT  tubes,  or  molded  shapes.  It  is  also  called  vulcanite  and  ebonite,  and  is  known  under 
various  trade  names.  It  is  a  hard,  dense  material,  is  easily  machinable,  takes  a  high  polish,  and  is 
resistant  to  wear.  At  temperatures  slightly  below  room  temperature  it  becomes  increasingly  brittle, 
and  at  higher  temperatures  it  softens  and  flows  under  pressure.  Under  heavy  load,  "cold  flow"  occurs 
at  room  temperature.  In  a  small  intermediate  temperature  range  it  is  tough  and  almost  "unbreak- 
able." It  is  combustible  but  is  not  easily  ignited.  It  has  low  water  absorption  and  is  immune  from, 
attack  by  most  acid  and  alkali  solutions  and  fumes.  It  is  attacked  'and  swelled  by  oils  and  rubber 
solvents.  It  is  attacked  by  ozone,  although  less  than  soft  rubber,  but  special  grades  are  available  with, 
improved  oil  and  ozone  resistance.  It  is  resistant  to  sparks  but  will  not  withstand  heavy  arcs.  The 
sulfur  is  never  fully  combined,  which  leads  to  some  serious  difficulties.  The  sulfur  has  a  tendency  to 
appear  in  a  surface  film  known  as  "bloom,"  causing  discoloration.  Ultraviolet  light  produces  sulfuric 
acid  from  this  layer,  seriously  lowering  the  surface  resistivity  and  causing  corrosion  of  nearby  metals. 
"Blooming"  can  be  greatly  reduced  by  careful  compounding.  Metallic  inserts  should  be  protected  by 
&  coating  of  tin  or  other  'corrosion-resistant  substance.  The  tendency  to  "cold  flow"  and  the  high  co- 

*  Trademark  names. 


2-44  PROPERTIES  OF  MATERIALS 

e^cient  of  expansion  can  be  reduced  by  incorporating  suitable  fillers  such  as  talc;  lower  grades  of  hard 
rubber  usually  are  filled  for  economic  reasons,  however.  Rubber  can  be  preformed  before  molding 
to  nearly  the  final  shape  and  hence  may  be  used  for  fine  tubes  and  thin-walled  sections  not  easily  ob- 
tainable with  phenolic  moldings,  but  the  accuracy  in  molding  is  usually  much  lower,  owing  to  shrinkage, 
distortion,  and  high  coefficient  of  expansion.  Hard  rubber  is  easily  machined  by  normal  methods,  but 
grinding  is  sometimes  more  economical.  Special  drills  also  give  improved  performance,  and  lubricants 
are  valuable  for  drilling,  tapping,  and  turning.  Tungsten  carbide  and  diamond  tools  give  more  nearly 
satisfactory  production.  The  material  may  be  sheared  and  punched  if  heated.  Many  moldings  as 
well  as  machined  parts  require  polishing  with  pumice  on  a  moderately  hard  "buff"  at  low  speed  to 
avoid  excessive  heat.  The  material  usually  is  black  but  can  be  obtained  in  a  number  of  colors,  mostly 
with  high  filler  content.  Hard  rubber  has  very  high  dielectric  strength  and  resistivity,  and  low  dielec- 
tric constant  and  power  factor,  but  all  the  dielectric  properties  are  affected  seriously  by  rising  tempera- 
ture. The  mechanical  temperature  limit  is  about  45  deg  cent  for  unloaded  and  70  deg  cent  for  loaded 
hard  rubber  with  light  pressures. 

Rubber,  Synthetic.  The  principal  synthetic  rubbers  are  Buna  S,  Buna  N,  neoprene,  butyl,  and 
Thiokol,*  although  elastomeric  vinyl  compounds  might  also  be  classed  as  synthetic  rubber.  All  the 
above  are  discussed  elsewhere  in  this  section. 

Rubber,  Vulcanized.    Sulfur  and  rubber  react  at  temperatures  in  the  vicinity  of  100  deg  cent  to 
form  a  tough,  elastic,  strong  material.    The  crude  rubber  is  washed,  sheeted,  and  dried.    The  sheets 
are  then  mixed  on  hot  rolls  with  sulfur,  fillers,  plasticizers,  accelerators,  and  anti-oxidants  as  wished, 
and  sheeted,  molded,  or  extruded  to  the  desired  forms.    The  article  then  is  vulcanized  by  heating  to 
temperatures  from  125  to  145  deg  cent  for  soft  rubber  articles,  and  160  to  170  deg  cent  for  hard  rubber 
parts.    With  plain  rubber,  from  2  to  10  per  cent  sulfur  gives  a  soft  rubber  stock.    Hard  rubber  contains 
from  20  to  32  per  cent  sulfur.    With  plain  sulfur,  vulcanization  or  "cure"  may  take  2  or  8  hours,  but 
by  means  of  accelerators  the  time  may  be  reduced  to  as  low  as  20  minutes.    Litharge,  lime,  and  mag- 
nesia are  inorganic  accelerators  as  well  as  fillers.    Complex  organic  compounds,  such  as  tetramethyl- 
thiuram  disulfide,  phenylguanidines,  and  mercaptobenzothiazole,  function  to  give  fast  cures  which  are 
not  critical  as  to  the  time  required  to  obtain  maximum  physical  properties  and  are  known  as  "flat" 
cures.    Fillers  in  the  form  of  fine  powders,  such  as  carbon  black,  zinc  oxide,  clay,  and  whiting,  are  em- 
ployed in  nearly  all  rubber  goods.    They  cheapen  the  compound,  of  course,  but  they  also  increase  the 
strength  and  toughness.  Soft  rubber  compounds  usually  contain  from  50  to  80  per  cent  of  filler.  Al- 
though the  natural  resins  and  proteins  assist  in  the  "milling"  or  breaking  down  of  the  rubber  to  some 
extent,  very  often  improved  working  is  obtained  by  adding  plasticizers  or  softeners,  including  paraffin, 
waxes,  para-cumaron  resin,  oils,  fats,  and  asphaltic  and  bituminous  materials  in  various  percentages. 
Reclaimed  rubber  also  improves  the  working  properties.    So-called  mineral  rubber,  which  is  an  asphaltic 
residue,  is  sometimes  added  up  to  20  per  cent  and  can  be  classed  as  a  filler.    The  dielectric  constant  of 
rubber-sulfur  compounds  at  25  deg  cent  rises  from  2  to  11  per  cent  of  sulfur  and  falls  again  from  16  to 
19  per  cent  sulfur,  and  then  changes  very  slightly  up  to  32  per  cent  sulfur.     The  power  factor  goes 
through  somewhat  similar  variation,  starting  at  8  per  cent  sulfur.    The  maxima  of  these  curves  are 
displaced  to  higher  sulfur  content  by  increasing  temperature.    The  normal  soft  and  hard  rubber  com- 
positions thus  have  low  dielectric  constant  and  power  factor,  and  the  intermediate  region,  which  is 
seldom  used,  has  poor  electrical  properties.    Resistivity  rises  in  a  fairly  regular  curve  from  2  to  28  per 
cent  sulfur.    (See  Bureau  of  Standards  Scientific  Paper  560,  part  II,  by  H.  L.  Curtis,  A.  T.  McPherson, 
and  A.  H.  Scott.)     Softeners  change  the  dielectric  constant  slightly  but  may  seriously  increase  the 
power  factor  in  quantities  of  only  10  per  cent.     The  dielectric  constant  is  increased  nearly  propor- 
tionally to  the  filler  content.    Carbon  black  causes  a  sharp  increase  of  dielectric  constant  from  2.7  to 
6.0.    Zinc  oxide,  lead  oxide,  and  selenium  show  much  slower  rates  of  increase.    Powdered  quartz  gives 
only  a  slight  increase.    The  effect  on  the  power  factor  is  much  the  same:  20  per  cent  carbon  black  ele- 
vates the  power  factor  from  0.0025  to  nearly  0.05.    Increasing  quartz  content  slightly  improves  the 
power  factor.    The  introduction  of  carbon  black  greatly  reducps  the  resistivity  of  rubber.     Carbon 
black  is,  however,  the  best  filler  from  a  mechanical  standpoint:  a  several-fold  increase  in  the  tensile 
strength  is  produced.    Rubber  compounds  absorb  water,  which  causes  an  increase  of  dielectric  con- 
stant and  power  factor,  and  decrease  of  resistivity,  but  on  long  immersion  the  power  factor  may  de- 
crease again.    Water  absorption  can  be  considerably  lowered  by  extended  washing  of  the  crude  rubber 
to  remove  water-soluble  matter  and  by  the  use  of  water-insoluble  fillers  such  as  silica,  zinc  oxide,  or 
hard  rubber  dust.    Absorption  of  water  is  less  in  sea  water  than  in  distilled  water.    The  usual  grade  of 
wire  insulation  contains  20  per  cent  minimum  of  rubber;  better  grades  have  30  per  cent,  and  high  grades 
40  per  cent.    Covering  of  portable  cords,  etc.,  subject  to  mechanical  wear  may  contain  up  to  60  per 
cent.    The  mechanical  and  dielectric  strength  of  rubber  is  lowered  by  the  action  of  oxygen  and  more 
rapidly  by  ozone  which  is  present  in  corona  discharge:  the  rubber  cracks  when  normally  vulcanized 
and  may  melt  if  the  temperature  is  high  and  the  compound  is  undervulcanized.    In  order  to  improve 
the  resistance  to  oxygen,  organic  compounds,  such  as  diphenylamines  or  hydroquinone,  known  as 
antioxidants,  are  added  in  small  percentage.    Rubber  stocks  are  tested  for  this  defect  under  300  Ib 
per  sq  in.  in  oxygen  gas  at  70  deg  cent  and  by  contact  with  ozone  at  atmospheric  pressure.    Stretching 
the  rubber  under  test  seriously  increases  the  rate  of  ozone  attack.    Ultraviolet  light  also  sharply  accel- 
erates the  combination  with  oxygen.    Heat  deteriorates  rubber  rapidly;  49  deg  cent  is  the  maximum 
operating  temperature  for  Code  rubber  insulation.    Performance  grades  are  satisfactory  at  60  deg  cent, 
and  heat-resistant  grades  at  75  deg  cent.   Rubber  is  swelled  quickly  and  eventually  dissolved  by  hydro- 
carbon solvents  and  oils.    Special  grades  are  available  which  minimize  this  defect.    At  low  tempera- 
tures a  low-sulfur  rubber  compound  is  no  longer  elastic,  and  a  piece  stretched  and  cooled  to  -20  dec 
cent  or  lower  will  not  return  to  its  original  length  until  the  temperature  rises.    The  power  factor  also 
increases  sharply  to  a  maximum  in  this  region,  with  a  value  over  10  times  the  value  at  20  deg  cent, 

*  Trademark  names. 


SOLID  DIELECTKIC  MATERIALS  2-45 

indicating  a  change  of  state  in  the  compound.    The  power  factor  also  rises  with  increasing  temperature, 
but  less  rapidly. 

Rutile  (TiO2>.    A  particular  crystalline  form  of  titanium  dioxide,  with  a  dielectric  constant  of  80 
to  110,  which  is  used  in  the  manufacture  of  high-dielectric-constant  ceramics. 
Saflex.*     Monsanto  Chemical  Co.     Polyvinyl  butyral. 
Saran.*     Dow  Chemical  Co.     Vinylidene  chloride. 

Saturated  Sleeving.  Cotton  sleeving  impregnated  with  thin  varnish  or  compound  which  does  not 
fill  completely  the  interstices  of  the  fabric.  Dielectric  strength  is  low,  and  resistance  to  humidity  is 
poor.  It  is  valuable  mainly  to  space  conductors  apart. 

Scotch  Tape.*  Minnesota  Mining  and  Mfg.  Co.  Pressure-sensitive,  non-corrosive  electrical  tape 
with  various  backing  materials. 

Shellac.  Produced  by  the  insect  Tachardia  lacca,  which  attaches  itself  to  numerous  species  of  trees 
native  to  India,  Indo-China,  and  Siam,  and  excretes  the  resin  at  several  parts  of  the  6-month  life  cycle. 
The  kusmi  tree  is  the  most  valuable  host  to  the  parasite  lac  insect  since  the  lac  therefrom  is  of  higher 
quality.  Crude  lac  consists  of  about  75  to  85  per  cent  resin,  2  to  4  per  cent  dye,  1  to  2  per  cent  ash, 
2  to  3  per  cent  water,  and  9  to  15  per  cent  residue  and  dirt.  The  natives  wash  the  crude  lac,  dry  it,  and 
press  it  through  a  bag  with  the  aid  of  heat.  The  mass  is  plastered  into  a  sheet  on  a  heated  object  and 
stretched  while  still  hot.  After  it  cools  it  is  broken  into  flakes  and  shipped  in  bags.  Native  shellac  is 
sometimes  adulterated  with  rosin.  Machine-made  shellac  is  extracted  by  various  processes  employing 
solvents  or  heat.  Refined  shellac  contains  over  90  per  cent  resin,  from  3  to  5  per  cent  of  wax,  1  to  2 
per  cent  moisture,  and  1  to  5  per  cent  of  matter  insoluble  in  alcohol.  Shellac  is  sold  in  many  grades 
which  are  principally  determined  by  the  color.  D.C.  is  a  very  high  grade,  free  from  dirt.  Superfine  is 
made  from  best  kusmi  lac.  Fine  and  Standard  No.  1,  T.N.,  and  garnet  lac  follow  in  descending  order 
of  quality.  Rosin  is  limited  by  trade  standards  to  3  per  cent  maximum  except  in  garnet  shellacs  which 
may  sometimes  contain  20  per  cent.  Machine-made  shellac  is  graded  somewhat  differently.  One 
manufacturer's  grades  are  Fine,  Superfine,  and  ABTN.  All  are  hard,  pure  shellacs.  BB  is  used  for 
blending,  and  amber  where  a  light  color  is  required.  Completing  the  list  are:  T.N.S.  pure  orange  shel- 
lac, various  grades  of  T.N.  shellac,  and  garnet.  Machine-made  garnet  is  low  in  wax  and  rosin,  and  is 
valuable  for  insulating  use.  Shellac  is  naturally  variable  in  quality,  depending  on  the  source  and 
methods  of  collection.  A  small  amount  of  orpiment  (arsenic  sulfide)  is  added  to  some  orange  shellac  to 
lighten  the  color.  Orpiment  is  insoluble  in  solvents,  but  otherwise  usually  has  no  beneficial  or  harmful 
effect.  Shellac  has  been  used  for  some  time  in  hot  molding  compounds,  in  insulating  varnishes,  and 
as  a  binder  for  composite  insulations  of  paper,  mica,  etc.  Various  mineral  fillers,  such  as  asbestos, 
powdered  mica,  clays,  marble,  and  wood  flours,  are  blended  with  shellac  on  rolls  in  a  manner  similar  to 
rubber  compounding.  The  material  is  sheeted  off  the  rolls,  and  blanks  are  cut  to  size.  The  preheated 
blanks  are  usually  molded  in  steam-heated  molds  at  about  160  deg  cent  under  hydraulic  pressure  for 
about  1  minute,  the  dies  are  then  chilled  with  water,  and  the  piece  is  removed.  Little  or  no  chemical 
action  takes  place.  The  finish  is  excellent  if  the  mold  surface  is  polished,  but  parts  can  be  polished  sub- 
sequently. Accuracy  of  molding  is  about  the  same  as  that  of  phenolic  moldings,  and  the  dies  are  much 
the  same  except  for  the  cooling  feature.  The  moldings  have  good  weather  resistance  and  are  fairly 
impervious  to  moisture,  but  they  are  somewhat  brittle  at  low  temperatures  and  soften  at  75  deg  cent. 
It  is  not  well  known  that  shellac  is  somewhat  thermosetting  under  certain  circumstances.  Long- 
continued  heating  above  100  deg  cent  will  solidify  the  melted  shellac  to  a  tough,  horny  mass.  This 
action  is  accelerated  by  increased  heat  and  by  hexamethylenetetramine,  aluminum  chloride,  urea,  and 
other  agents,  and  is  retarded  by  alkalies,  alkaline  salts,  aniline,  and  other  substances.  (See  Bulletin  14, 
Indian  Lac  Research  Institute.)  Shellac  is  soluble  in  alcohols  and  ketones,  and  the  solutions  are  used 
for  varnishes  and  cements.  Shellac  loses  solubility  on  standing  for  long  periods  of  time,  and  the  plas- 
ticity is  also  decreased  somewhat.  The  flexibility  of  shellac  films  may  be  increased  by  plasticizing  with 
castor  oil  or  tricresyl  phosphate. 

Silaneal.*     Dow  Corning  Corp.     Silicone  treating  fluid  for  ceramics. 
Silastic.*    Dow  Corning  Corp.     Silicone  rubbers. 

Silica,  Fused.  Fused  silica  is  similar  to  translucent  fused  quartz  in  its  properties,  but  it  is  not  made 
from  as  pure  a  sand  and  usually  contains  some  iron.  It  is  an  excellent  insulator. 

Silicones.  Silicones  are  a  class  of  organo-silicon  compounds  with  a  chemical  structure  analogous 
to  that  of  hydrocarbons,  but  with  the  carbon  atoms  of  the  chain  replaced  by  silicon  atoms  with  an 
oxygen  atom  inserted  in  each  bond  between  the  silicon  atoms.  By  attaching  various  hydrocarbon 
chains  to  each  silicon  atom,  and  varying  the  chain  length  by  different  polymerization  procedures,  fluids, 
greases,  plastics,  and  resins  are  produced.  The  general  properties  of  silicones  compared  to  those  of 
hydrocarbons  are:  (1)  improved  heat  resistance;  (2)  smaller  change  in  viscosity  with  temperature; 
(3)  resistance  to  oxidation;  (4)  resistance  to  arcing;  (5)  high  flash  and  fire  points.  In  common  with 
some  hydrocarbons,  they  have  low  power  factors  and  are  water-repellent  to  a  high  degree.  Silicone 
fluids  are  available  in  viscosities  from  0.65  to  1000  centistokes  at  25  deg  cent,  and  in  volatilities  from 
nearly  zero  to  approximately  that  of  water.  They  have  dielectric  constants  of  2.4  to  2.75,  and  power 
factors  of  0.0002  from  100  cycles  to  10  Me,  rising  to  0.0006  at  100  Me.  Power  factor  increases  with 
temperature  but  is  always  less  than  that  of  a  good  grade  of  mineral  oil.  Silicone  fluids  may  be  used  to 
treat  gla^ss  or  ceramics  (Silaneal;*  Dri-film  *)  to  produce  a  gwater-repellent  surface,  giving  a  greatly 
increased  surface  resistivity  under  condensation  conditions.  Silicone  greases  (DC  No,  4  Ignition  Seal- 
ing Compound)  may  be  used  to  fill  connectors  to  prevent  arc-over,  corona,  or  leakage,  or  to  render 
surfaces  water-repellent.  Silicone  resins  are  used  in  conjunction  with  glass  textiles  to  form  heat- 
resistant  boards,  cloths,  and  wire  insulation,  and  in  the  form  of  varnishes  for  coil  impregnation.  Sili- 
cone rubber  (Silastic  *)  is  heat-resistant  to  250  deg  cent,  remains  flexible  down  to  —55  deg  cent,  and 
has  good  dielectric  properties. 

*  Trademark  names. 


2-46  PROPERTIES  OF  MATERIALS 

Bass,  S.  L.f  and  T.  A.  Kaupp,  Proc.  I.R.S.<  Vol.  33,  441  (July  1945). 

Johannson,  O.  K,  and  J.  J.  Torok,  Proc.  I.R.E.,  Waves  and  Electrons  section,  Vol.  34,  296  (May 
1946). 

Norton,  F.  J.,  Gen.  Elec.  Rev.,  August  1944. 

Silk.  Silk  is  obtained  from  cocoons  spun  from  double  continuous  filaments  secreted  by  the  "silk- 
worm," which  is  the  larva  of  the  Bombyx  mori  and  other  moths.  The  fiber  is  unwound  from  the  cocoon 
by  hand,  usually  scoured  (degummed)  to  remove  the  natural  sticky  gum  or  wax,  called  sericin,  which 
cements  the  duplex  filaments,  and  twisted  into  thread.  Cultivated  silkworms  are  fed  on  mulberry 
leaves;  wild  silkworms  give  a  coarser  quality  known  as  tussah  silk.  Orgazine  silk  is  from  the  best 
selected  cocoons,  and  tram  silk  is  from  the  poorer  cocoons.  Floss  silk  is  spun  from  broken  lengths  of 
filaments.  Silk  for  insulation  should  be  free  of  loading  materials  and  as  well  washed  as^  possible,  since 
thorough 
plain  or  v 

strength  ^  UJt — ^..   _„„_  _ „_ __...„. 

constant  but  is  not  as  resistant  to  heat.     Silk  flock  is  sometimes  used  to  add  strength  to  molding 
compounds. 

Sisal  Hemp.  A  bast  cordage  fiber  obtained  from  the  leaves  of  the  century  plant  or  agave.  In 
strength  and  length  of  fiber  it  is  inferior  to  manila  hemp.  It  is  used  to  some  extent  in  making  paper 
and  pressboard,  and  for  reinforcing  large  molded  laminated  parts. 

Spauldite.*     Spaulding  Fibre  Co.     Phenolic  laminates. 

Steatite.  Principal  ceramic  used  for  radio  apparatus.  Made  chiefly  from  magnesium  silicate 
which,  after  firing,  forms  clinoenstatite  crystals.  Low  water  absorption  and  excellent  electrical  prop- 
erties are  characteristic.  Special  grades  (L-5)  are  available  with  even  lower  losses  than  standard  or 
regular  grades  (L-4  or  L-3).  Specifications:  Joint  Army-Navy  Spec.  JAN-I-10,  Grades  L-3,  L-4,  or  L-5. 

Styraloy.*     Dow  Chemical  Co.     Elastomeric  polystyrene  copolymer. 

Styramic.*     Monsanto  Chemical  Co.    Polystyrene  and  chlorinated  diphenyl  molding  compound. 

Styramic  H.  T.*    Monsanto  Chemical  Co.    Polydichlorostyrene  molding  compound. 

Styrene.  Volatile  liquid  monomer,  also  known  as  vinyl  benzene,  used  for  manufacture  of  poly- 
styrene, Buna  S  rubber,  and  other  plastics.  Polymerizes  spontaneously  in  time  to  polystyrene  or 
more  rapidly  with  the  aid  of  heat  or  a  catalyst.  It  also  is  used  as  a  fully  reactive  constituent  in  poly- 
ester laminating  liquids  so  that  no  solvent  evaporation  is  necessary. 

Styrofoam.*    Dow  Chemical  Co.     Expanded  polystyrene. 

Styron.*    Dow  Chemical  Co.     Polystyrene  of  various  types. 

Synthane.*     Synthane  Corp.     Phenolic  laminates. 

Teflon.*    E.  I.  du  Pont  de  Nemours  &  Co.     Polytetrafltioroethylene  products, 

Tegit*     Garfield  Mfg.  Co.    Asbestos  coal-tar  moldings. 

Tenite  I.*     Tennessee  Eastman  Corp.     Cellulose  acetate. 

Tenite  IE.*     Tennessee  Eastman  Corp.     Cellulose  acetate-butyrate. 

Textolite.*    General  Electric  Co.    Phenolic  and  other  molded  or  laminated  products. 

Thalid.*    Monsanto  Chemical  Co.     Low-pressure  or  contact  laminating  resins. 

Thiofcol.*    Thiokol  Corp.    Polysulfide  rubbers. 

Transite.*     Johns-Manville  Corp.     Portland  cement  and  asbestos  molded  products. 

Tnf-nex.*     Libby-Owens-Ford  Glass  Co.     Tempered  glass. 

Tun.*     International  Products  Corp.     Glass-bonded  mica. 

Urea  Resins.  Reaction  of  urea,  CO(NH2)2,  and  formaldehyde  produces  methylol  ureas  which  are 
water-soluble.  Paper,  alphacellulose,  cloth,  or  wood  is  impregnated  with  solutions  and  cured  with 
heat  and  catalysts  to  a  thermosetting,  water-insoluble  plastic.  Urea  moldings  are  light  weight,  rigid, 
and  hard.  They  have  high  dielectric  strength,  good  arc  resistance,  and  moderate  electrical  losses. 
Impact  strength  is  lower  than  that  of  phenolic  materials.  Translucent  moldings  in  any  color  may  be 
obtained.  Specifications:  ASTM  D705.  Trade  names:  Beetle,  Plaskon. 

Varnishes,  Insulating.  Varnishes  are  generally  classified  according  to  composition,  as  oleoresinous 
or  4'oil  varnishes"  and  "spirit  varnishes,"  but  some  commercial  types  do  not  fall  strictly  into  either 
class.  Oil  varnishes  are  made  by  combining  a  resin,  commonly  a  copal,  with  a  drying  oil.  It  is  usually 
necessary  to  melt  or  "run"  the  resin  before  adding  the  oil  to  get  a  clear  solution.  Varnishes  with  a 
high  oil  content  are  known  as  "long-oil"  varnishes;  they  are  slow  drying  but  deposit  very  flexible  films. 
"Short-oil"  varnishes  have  high  resin  content  and  deposit  a  hard  film.  Oil  varnishes  harden  as  a  result 
of  oxidation  and  polymerization  of  the  drying  oils,  such  as  linseed,  tung  (China  wood) ,  or  soya  bean, 
and  often  of  the  resin  or  asphalt  as  well.  Drying  is  accelerated  by  adding  small  percentages  of  cata- 
lytic agents,  called  "driers,"  usually  in  the  form  of  resinates  or  linoleates  of  cobalt,  lead,  or  manganese, 
which  increase  the  rate  of  oxidation.  Short-oil  varnishes  will  air-dry  in  4  to  18  hr,  depending  on  the 
type,  to  a  reasonable  degree  of  hardness.  Long-oil  varnishes  will  not  air-dry  in  a  reasonable  time.  By 
baking  at  100  to  110  deg  cent  the  drying  time  can  be  shortened  to  2  to  8  hr  because  of  the  faster  oxida- 
tion. Baking  produces  a  harder  film  and  gives  better  adhesion  to  the  object.  It  also  serves  to  drive 
out  moisture  from  fibrous  materials  which  are  being  treated,  and  to  improve  the  dielectric  properties. 
Many  modern  insulating  varnishes  contain  synthetic  resins  of  the  phenol-aldehyde  or  alkyd  type  which 
give  hard  durable  films  as  the  result  of  the  thermosetting  of  the  resins  during  baking.  Thermosetting 
resins  are  also  used  dissolved  in  "spirit"-type  solvents  instead  of  oils,  yielding  a  moderately  nard  film 
on  air-drying,  which  is  increased  in  hardness  and  durability  by  baking.  The  dielectric  properties  of 
the  synthetic  resin  varnishes  are  usually  excellent.  Asphalts  are  used  with  resins  in  some  black  oil 
varnishes  and  without  oil  or  resin  in  the  so-called  asphaltum  varnishes  which  are  merely  solutions  of 
asphalts  in  benzine  or  other  hydrocarbons.  These  varnishes  dry  principally  by  evaporation,  but  the 
last  traces  of  solvent  leave  the  asphalt  very  slowly,  and  if  the  object  is  heated  to  drive  off  the  solvent 
many  of  the  asphalts  will  oxidize  and  polymerize  to  a  certain  extent,  producing  a  fairly  hard  film. 

*  Trademark  names. 


SOLID  DIELECTRIC  MATERIALS 


2-47 


Spirit  varnishes  dry  by  evaporation  of  the  solvent,  leaving  a  film  of  the  dissolved  resin,  asphalt,  or 
gum.  Cellulose  nitrate  and  acetate  varnishes  are  in  this  class  but  are  usually  considered  separately 
as  "lacquers."  Many  of  the  resins  leave  a  brittle  film  when  used  alone  so  that  a  soft  gum  or  plasticiz- 
ing  agent  like  castor  oil  is  usually  added  to  give  the  necessary  flexibility,  except  in  shellac  which  ordi- 
narily does  not  need  plasticizing.  The  principal  spirit  varnish  resins  are  shellac,  manila  copal,  dammar, 
mastic,  kauri,  and  sandarac.  The  solvents  and  thinners  used  are  alcohols,  esters,  hydrocarbons,  and 
turpentine.  Spirit  varnishes  are  commonly  air-drying,  although  the  evaporation  of  the  solvent  is  often, 
hastened  by  moderate  heating.  Spirit  varnishes  are  generally  not  used  for  impregnation  but  are  com- 
mon for  external  coats  and  for  sticking,  and  for  bonding  of  mica  and  other  materials.  For  external 
coating  work,  varnishes  are  applied  by  spraying,  dipping,  roller  coating,  or  brushing,  depending  on  the 
nature  of  the  work.  For  impregnation,  dried  articles  are  dipped  while  still  hot  into  a  varnish  of  low 
viscosity  and  low  surface  tension  to  insure  thorough  penetration.  Dipping  time  must  be  determined  for 
each  article  by  experiment.  A  much  better  impregnation  is  secured  by  drying  coils  in  a  vacuum  and  ad- 
mitting the  varnish  to  the  work  container,  then  "breaking"  the  vacuum  and  applying  pressure.  Impreg- 
nated coils  should  be  drained  and  baked  at  100  to  110  deg  cent  for  a  period  sufficient  to  harden  the 
varnish  film.  Higher  temperature  tends  to  disintegrate  fibrous  materials  in  prolonged  baking,  and 
lower  temperatures  do  not  remove  moisture.  For  short  drying  schedules,  temperatures  up  to  150  deg 
cent  are  sometimes  employed  satisfactorily.  Objects  should  be  exposed  to  fresh  currents  of  air  during 
drying  in  order  to  remove  solvent  vapors  which  retard  hardening.  For  methods  of  testing  dielectric 
strength,  heat  endurance,  and  oil  proofness,  see  ASTM  D115. 

Varnished  Cloth.  A  suitable  fabric  coated  with  yellow  or  black  insulating  varnish  so  that  the 
fibers  are  thoroughly  impregnated.  Varnished  silk  usually  is  from  0.003  to  0.005  in.  and  cotton  from 
0.005  to  0.040  in.  thick.  Tensile  strength  of  cotton-base  cloth  per  inch  width  of  warp  runs  from  45  to 
100  Ib,  and  Elmendorf  tearing  strength  across  the  warp  varies  from  100  to  300  grams.  Dielectric 
strength  usually  runs  from  800  to  1500  volts  per  mil. 

Varnished  Tubing.  Commonly  called  "spaghetti,"  magneto  tubing,  etc.,  according  to  manufacturer 
and  grade;  made  by  coating,  or  impregnating  and  coating,  cotton  sleeving  with  varnishes.  ASTM 
Specification  D372  distinguishes  three  grades  (see  specification  for  details).  Grade  A  is  generally  known 
as  flexible  varnished  tubing,  or  motor  and  transformer  tubing,  or  impregnated  magneto  tubing;  it  has 
a  minimum  dielectric  strength  of  7000  volts  average.  Grade  B,  generally  known  as  "radio  spaghetti," 
has  a  minimum  average  dielectric  strength  of  4000  volts.  Grade  C  is  similar  to  saturated  sleeving,  and 
its  dielectric  strength  is  lower. 

Vibrin.*  Naugatuck  Chemical  Co.,  Div.  U.  S.  Rubber  Co.  Liquid  polyesters  and  cross-linking 
monomers. 

Vinyl  Plastics.     An  extremely  important  class  of  thermoplastic  resins,  composed  of  linear  chains 


formed  by  the  polymerization  of  monomers  of  the  general  type 


tinuous  chain 


If  group  X  is 

hydrogen 

hydrogen 

chlorine 

hydrogen 

hydrogen 

hydrogen 

hydrogen 


The  result  is  a  con- 


and group  Y  is 


the  product  is 


hydrogen 

chlorine 

chlorine 

hydroxy  ( — OH) 

phenyl  (CeHs— ) 

carbazyl  (C^HsN — ) 

acetoxy  (CH3-CO-0— ) 


polyethylene  (Polythene  *) 

polyvinyl  chloride  (Geon  *) 

polyvinylidene  chloride  (Saran  *) 

polyvinyl  alcohol 

polystyrene 

polyvinyl  carbazole  (Polectron  *) 

polyvinyl  acetate 


methyl  (CH3)        methylcarboxy  (CH30-OC— )       methyl  methacrylate  (Plexiglas*  Lucite*) 

If  mixtures  of  two  monomers  are  copolymerized,  a  copolymer  such  as  polyvinyl  chloride-acetate  is 
produced  in  which  all  the  Y's  are  hydrogen,  most  of  the  X's  are  chlorine,  and  the  remainder  of  the  X's 
are  the  acetoxy  group.  The  polyvinyl  formals  are  a  general  group  of  polymers  with  modified  chains 
constructed  like  this:  (etc.— CH2— CH— CHs— CH-etc.) 

A       A 


If  group  Z  J8 

hydrogen 
methyl  (CH3— ) 
butyl  (C3Hr- ) 


the  product  is 


polyvinyl  formal  (Formex,*  Formvar  *) 

polyvinyl  acetal  (Alvar  *) 

polyvinyl  butyral  (Butacite*  Butvar^*  Saflex,*  Vinylite*  X) 


Of  course,  different  conditions  of  temperature,  pressure,  catalyst,  and  carrier  solvent  or  emulsion,  are 
required  for  each  case,  and  the  length  of  the  chain  of  molecules  is  varied  to  suit  requirements  by  altering 
these  conditions.  Only  those  plastics  with  vinyl  in  the  product  name  are  usually  classed  as  vinyl  plas- 
tics, but  by  this  illustration  the  relationship  of  many  of  the  thermoplastics  is  immediately  apparent. 

Vinyl  Chloride.     Perhaps  the  most  important  vinyl  plastic  from  a  tonnage  viewpoint,  vinyl  chloride 
is  very  extensively  used  for  hook-up  wire  insulation,  cable  jackets,  insulating  tubing,  and  tape.    It  is 

*  Trademark  names . 


2-48  PROPERTIES   OF  MATERIALS 

also  employed  for  non-rigid  or  elastomeric  molded  parts  since  it  has  rubberlike  characteristics  when 


of  plasticizer,  but,  in  general,  these  resins  have  a  rather  high  dielectric  loss  and  a  lower  insulation  re- 
sistance than  those  of  rubber  compounds,  although  they  have  a  high  dielectric  strength.  Wire  insulation 
for  operating  temperatures  of  80  deg  cent  is  now  available,  although  60  deg  cent  has  been  the  limiting 
temperature  for  most  older  compositions.  These  compounds  are  nearly  non-inflammable  and  do  not 
oxidize  like  rubber  but  are  subject  to  some  stiffening  due  to  loss  of  plasticizer  with  time.  Except  for 
flexing  at  low  temperature  the  life  should  be  very  long.  Specifications  for  resin:  ASTM  D72S.  Trade 
names:  Flamenol,  Geon  100  series,  Vinylite  Q  series,  Koroseal. 

Vinyl  Chloride-acetate.  This  resin  is  similar  to  straight  polyvinyl  chloride  resin  except  that  about 
5  per  cent  of  the  resin  may  be  vinyl  acetate,  which  improves  the  processing  and  extrusion  characteristics 
and  reduces  the  amount  of  plasticizer  required.  "Uses  and  characteristics  are  similar  to  those  of  poly- 
vinyl  chloride,  but  in  addition  these  resins  are  used  for  rigid,  transparent,  or  colored  sheets  of  high 
dimensional  stability  and  good  electrical  properties.  These  resins  also  have  proved  to  be  very  success- 
ful for  high-quality  phonograph  records  and  for  manufacture  of  textiles  (Vinyori).  Specifications  for 
rigid  sheet:  ASTM  D708;  for  non-rigid  resin:  ASTM  D742.  Trade  names-  Vinylite  V  series. 

Vinylidene  Chloride.  Similar  to  vinyl  chloride  resins  but  contains  more  chlorine,  is  harder,  and  is 
capable  of  orientation  by  drawing  or  other  cold  work  to  increase  the  tensile  strength.  ^For  molding 
purposes,  it  is  commonly  copolymerized  with  10  per  cent  of  vinyl  chloride  to  serve  as  an  internal  plas- 
ticizer. Though  the  electrical  properties  are  not  outstanding,  the  strength,  toughness,  and  non-inflam- 
mability of  this  plastic  have  made  it  valuable  for  fungus-resistant  cable  braids  and  ropes,  as  tubing  for 
water  cooling,  etc.  Specifications  for  molding  compounds:  ASTM  D729.  Trade  name:  Saran. 

Vinylite.*  Carbide  and  Carbon  Chemical  Co.  Vinyl  resins,  of  which  various  grades  are  distin- 
guished by  suffix  letters. 

Vinyon.*    Carbide  and  Carbon  Chemical  Co.    Polyvinyl  chloride-acetate  textiles. 

Vistanex.*     Standard  Oil  Co.  of  N.  J.     Polyisobutylene. 

Vitreosil.*    Thermal  Syndicate,  Inc.    Fused  silica. 

Voltron.*    Industrial  Synthetics  Corp.    Vinyl  tubing  and  tape. 

Vulcabeston.*     Johns-Manville  Corp.     Asbestos  with  rubber  or  gum  binder. 

Vulcoid.*     Continental-Diamond  Fibre  Co.     Resin-impregnated  vulcanized  fiber. 

Vycor.*     Corning  Glass  Works.     High-silica  glass* 

Waxes.  Waxes  are  of  three  origins:  animal,  vegetable,  or  mineral.  The  principal  animal  waxes 
are:  beeswax,  m.p.  63  deg  cent;  wool  wax,  m.p.  35  deg  cent;  spermaceti,  m.p.  49  deg  cent;  insect  or 
Chinese  wax,  m.p.  81  deg  cent.  The  principal  vegetable  waxes  are:  carnauba,  m.p.  85  deg  cent,  and 
candelilla,  m.p.  68  deg  cent.  The  principal  mineral  waxes  are:  montan,  m.p.  72  deg  cent;  ozokerite 
refined  to  form  ceresin,  m.p.  65-73  deg  cent;  paraffin  waxes  obtained  from  petroleum.  Waxes  in 
general  are  not  "wetted"  by  water  and  are  very  resistant  to  penetration  by  moisture.  Ceresin  and 
non-crystallizable  high-melting-point  paraffin  waxes  such  as  Superla,*  Syncera,*  and  Cerese  wax  *  have 
higher  insulation  and  moisture  resistance  than  the  low-melting  paraffins  and  the  animal  and  vegetable 
waxes.  Waxes  which  crystallize,  such  as  montan,  have  a  tendency  to  crack  and  admit  moisture  by 
capillary  action;  mixture  with  a  soft  wax  or  resin  usually  minimizes  this  tendency.  Wax  compounds 
are  hardened  and  the  flow  point  raised  by  incorporating  rosin  or  other  resins  and  higher-melting  waxes 
such  as  ceresin,  montan,  or  carnauba.  Most  naturally  occurring  waxes  have  low  dielectric  constants 
and  low  power  factors  as  well  as  high  resistivity.  Synthetic  waxes  made  by  chlorinating  naphthalene 
or  paraffin  have  somewhat  higher  dielectric  constants  and  losses.  Long  heating  of  liquid  waxes  at 
high  temperatures  in  contact  with  air  tends  to  cause  decomposition  and  development  of  acidity,  par- 
ticularly in  the  presence  of  copper,  which  acts  as  a  catalytic  agent  to  increase  oxidation  markedly. 

Zircon  Porcelain.  This  material  consists  chiefly  of  zirconium  silicate  with  addition  of  small  amounts 
of  clay  and  metallic  oxides  to  aid  in  manufacture.  It  has  higher  mechanical  strength  than  other 
electrical  ceramics,  and  a  lower  coefficient  of  expansion  than  all  but  cordierite.  It  is  extremely  re- 
sistant to  thermal  shock.  The  electrical  characteristics  at  radio  frequencies  are  excellent,  except  that 
the  dielectric  constant  is  50  per  cent  higher  than  that  of  steatite  and  changes  more  rapidly  with  tem- 
perature. The  resistivity  at  high  temperatures  is  about  the  same  as  that  of  steatite  and  is  superior 
to  that  of  cordierite  and  high-voltage  porcelain. 

6.  LIQUID  DIELECTRICS 

Dielectric  Constant.  The  dielectric  constant  of  liquids  ranges  from  about  1.5  to  almost 
100.  Non-polar  liquids  at  room  temperature  have  constants  of  1.84  to  2.3,  which  are 
nearly  equal  to  the  square  of  the  index  of  refraction.  The  dielectric  constants  of  non-polar 
liquids  are  independent  of  frequency  and  vary  only  slightly  with  temperature,  owing  to 
thermal  expansion.  On  the  other  hand,  polar  liquids  have  higher  dielectric  constants 
which  vary  to  a  marked  degree  with  temperature  and  frequency  in  certain  regions  When 
a  polar  liquid  freezes,  part  of  the  influence  of  the  dipoles  is  lost  and  the  dielectric  constant 
drops  very  sharply.  High  rates  of  change  in  dielectric  constant  with  frequency  or  tempera- 
ture are  normally  associated  with  high  power  factors  at  the  same  frequency  and  tempera- 
ture.  The  dielectric  constants  and  dipole  moments  of  some  liquids  are  shown  in  Table  2. 

*  Trademark  names. 


LIQUID  DIELECTRICS 


2-49 


Table  2.    Dielectric  Properties  of  Liquids 


Substance 

Temper- 
ature, 
deg 
cent 

Wave- 
length , 
cm 

Dielectric 
Constant 
(X) 

Temper- 
ature 
Coeffi- 
cient of  K, 

io~4/°c 

(negative) 

Polar 
Moment, 

io-18 

esu 

Conductivity, 
mhos  /cm 

Acetone  

-80 

oo 

33.8 

0 

oo 

26  6 

« 

+  15 

1200 

21   85 

a 

17 

73 

20  7 

a 

25 

00 

21   3 

2  75 

6  X   10~8 

Air  (liquid)  

-  191 

00 

1   43 

0 

Amyl  acetate                       .  • 

19 

oo 

4  81 

24 

1   91 

\rnyl  alcohol            

20 

00 

16  0 

1   7 

18 

200 

10  8 

«           tt 

18 

73 

4  7 

Aniline          

18 

00 

7.32 

35 

1   56 

2  4  X  10~~8 

Benzene 

20 

00 

2  282 

8  6 

o 

7  6  X  10~~18 

7i-Butyl  alcohol 

20 

oo 

17  4 

76 

1   65 

Butyl  stearate  

25 

oo 

3.3 

Carbon  dioxide  (liquid)  
Carbon  tetrachloride 

-5 
20 

00 

00 

1.60 
2  3 

12  5 

0 
0 

4  X  10~18 

Castor  oil  

00 

4.67 

107 

+ 

1    7  x  10~u 

Chlorinated  diphenyl 
mobile            .    .        

25 

5.8 

163 

viscous     

25 

oo 

5.05 

133 

Chlorobenzene 

20 

oo 

5  72 

31   0 

1   56 

Chloroform.       

20 

00 

4.84 

39  0 

1    1 

<2  X  10~8 

Cumene  

22 

00 

2.2 

0  4 

Cyclohexane 

20 

00 

2  41 

8  6 

o 

Cymene           

22 

00 

2  5 

? 

<2  X  10~8 

Decahydronaphthfliene 

20 

00 

2  26 

6  6 

o 

Decane  

20 

00 

1.991 

6.7 

0 

Decylene 

17 

oo 

2  21  1 

o-Dichlorobenzene  
wi-Dichlorobenzene          . 

20 
20 

oo 

OO 

10.2 
5   11 

45 
28  0 

2.25 
1   48 

- 

Dodecane                

20 

00 

2  017 

7  4 

o 

Ethyl  abietate  

20 

09 

3.95 

ca.  20 

+ 

Ethyl  acetate 

20 

6   15 

1   85 

<  1  X  1  0~  9 

Ethyl  alcohol                    .    . 

Frozen 

00 

2  7 

-120 

00 

54.6 

u            it 

-80 

oo 

44  3 

ti           u 

—  40 

00 

35  3 

u                st 

0 

00 

28.4 

u                u 

+20 

OO 

25  8 

63  0 

1   68 

1   3  x  10~9 

it                u 

17 

200 

24.4 

U                       il 

17 

75 

23  0 

U                      It 

17 

53 

20.6 

it          It 

17 

*      4 

8  8 

u               tt 

17 

0.4 

5  0 

Ethyl  benzene 

22 

00 

2  2 

0  5 

Ethyl  ether  

20 

00 

4.4 

46 

1    15 

<4  X  10~18 

Ethylene  glycol 

20 

00 

38  8 

2  28 

3  X  IO"7 

Glycerin              ... 

25 

oo 

43  0 

ca   52 

+ 

6  4  X  10~"8 

15 

1200 

56.2 

+ 

u 

15 

200 

39   i 

te 

15 

75 

25  4 

tc 

8.5 

4.4 

_ 

1C 

0  4 

2  6 

Heptane     

20 

00 

1  926 

8  6 

o 

4  X  10~13 

Hexane  ... 

20 

00 

1   890 

9  0 

o 

4  X  10~18 

Hydrog^Tx  (1  in  uid) 

—  258  4 

00 

1   241 

o 

"Kerosene   .                     , 

25 

ca   2   1 

? 

<1   7  X  10~8 

3Vtesitylene 

22 

oo 

2  2 

o 

Methyl  alcohol 

Frozen 

00 

3  07 

—  100 

00 

58  0 

«             « 

—  50 

00 

45  3 

«            « 

0 

oo 

35.0 

_ 

tt            n 

+20 

00 

31.2 

57 

1   68 

4  4  X  10~~7 

Mineral  oil  

20 

00 

2.  191 

4.7 

0 

1  X  10~16 

2-50 


PROPERTIES  OF  MATERIALS 


Table  2.    Dielectric  Properties  of  Liquids — Continued 


Substance 

Temper- 
ature, 
deg 
cent 

Wave- 
length, 
cm 

Dielectric 
Constant 

TO 

Temper- 
ature 
Coeffi- 
cient of  K, 

icrV0c 

(negative) 

Polar 
Moment, 

io-18 

esu 

Conductivity, 
mhos/cm 

Nitrobenzene 

-30 

00 

3.1 

— 

_ 

u 

-13 

co 

3.2 

— 

— 

a 

^ 

CO 

3.4 

— 

— 

u 

-4 

00 

3.8 

— 

— 

u 

+  15 

00 

37.8 

— 

— 

u 

18 

co 

36.45 

3.9 

2  X  10^8 

u 

30 

00 

35.1 

_ 

— 

Nitrogen  Qiquid)                    » 

-208 

00 

1.44 

_ 

0 

— 

Octane         .                

20 

00 

1.949 

7.6 

0 

— 

20 

00 

3.11 

36 

_ 

2  X  10~13 

Peanut  oil   .                             « 

11.4 

00 

3.03 

— 

— 

Pentane       .               , 

20 

00 

1.845 

9.6 

0 

<2  X  10~10 

Petroleum 

2000 

2.13 

_ 

_ 

3  X  10~13 

Petroleum  etner             .  . 

20 

00 

1.92 

_ 

0 

— 

Phenol   

45 

oo 

10.3 

_ 

1.73 

<  1  .  7  X  1  0~8 

•w-Propyl  alcohol 

20 

00 

22.2 

53 

1.66 

5  X  IO-8 

Pyridine      .      ....      .*.... 

22 

00 

13.9 

2.1 

5.3  X  10~8 

Quinoline  

22 

00 

9.0 

_ 

2.25 

2.2  X  10~8 

Rosin  oil   

20 

00 

2.55-2.8 

ca.  22 

— 

— 

Silicone  fluids 
DC  200t  200  centistokes.. 
DC  500,  20  eentistokes... 
Toluene          

20 
20 
-83 

00 

oo 
oo 

2.76 
2.71 
2.51 

34 
31 

0 
0 

,0-H 
]0-14 

+  16 

00 

2.33 

_ 

_ 

_ 

it 

19 

73 

2.31 

9.8 

0.52 

<i  x  io-14 

Turpentine 

20 

00 

2.23 

2  X  10~13 

18 

00 

2.376 

8.2 

0.4 

<1  X  10~15 

17 

73 

2.37 

_ 

p-Xylene      

20 

00 

2.25 



0 

_ 

Water  (pure),  frozen  , 

—  18 

5000 

3.16 

_ 

_ 

_ 

ij 

1200 

2.85 





1.6X  10~* 

liquid 

+  17 

200 

80.6 



17 

74 

81.7 







u 

17 

38 

83.6 

_ 

a 

18 

00 

81.07 





4  X  10~8 

« 

50 

_ 

- 

1.7X  10~7 

Notes.    A  wavelength  greater  than  10,000  cm  is  denoted  by  °° . 
Zero  indicates  that  substance  has  no  polar  moment. 
Plus  sign  indicates  that  substance  is  polar  but  value  for  polar  moment  was  not  available. 

D-c  Conductivity  and  Resistivity.  The  d-c  conductivity  of  liquids  is  ionic  in  nature 
and  has  a  high  positive  temperature  coefficient.  Change  in  conductivity  with  temperature 
is  expressed  by 

0  =  G0ea/T     or    G  =  GQebt 

where  £0,  a,  and  &  are  constants,  T  is  the  absolute  temperature,  and  t  is  the  temperature  in 
degrees  centigrade  for  small  temperature  differences.  Thus  a  plot  of  the  logarithm  of 
either  conductance  or  resistivity  against  I/ T  is  a  straight  line,  and  against  t  is  approxi- 
mately straight  for  small  temperature  intervals.  The  increase  in  conductivity  with 
temperature  is  the  result  of  an  increase  in  ionic  mobility  arising  from  the  reduction  in 
viscosity.  Log  resistance-temperature  curves  therefore  change  slope  at  regions  where  the 
viscosity  varies  sharply,  as  at  freezing  or  transition  points. 

The  conductivity  of  pure  liquids  may  be  increased  enormously  by  small  amounts  of 
impurities  or  moisture  which  readily  ionize  in  the  particular  liquid.  Fortunately  the 
degree  of  ionization  is  a  function  of  the  dielectric  constant,  so  that  the  non-polar  liquids 
having  a  low  dielectric  constant  are  less  sensitive  to  impurities,  especially  in  low  concen- 
tration. The  difficulty  of  preventing  contamination  of  liquids  of  higher  dielectric  constant 
has  effectively  prevented  their  use  for  capacitors.  The  resistivities  characteristic  of  com- 
mercially pure  liquids  are  approximately  in  inverse  relationship  to  the  dielectric  constants, 


LIQUID  DIELECTRICS 


2-51 


as  is  shown  by  the  following  table  taken  from  The  Properties  of  Dielectrics,  by  F.  M. 
Clark,  in  the  J.  Franklin  Inst.t  Vol.  208,  17  (July  1929). 

Table  3.    Relation  between  Dielectric  Constant  and  Resistivity 


Material 

Dielec- 
tric 
Con- 
stant 

Characteristic 
Resistivity, 
ohm-cm 

Material 

Dielec- 
tric 
Con- 
stant 

Characteristic 
Resistivity, 
ohm-cm 

Benzene  
Petroleum  oils  .  . 

2.15 
2.2 

4.7    X  1012 
10.0    X  1012  (100°  C) 

China  wood  oil  ... 
Castor  oil  

3.5 
4.3 

0.08  X  1012  (100°  C) 
0.06  X  1012         " 

Paraffin  wax  .  .  . 
Cottonseed  oil.  . 
Asphalt  

2.25 
2.9 
3.  1 

5.0    X  I012 
0.2    X  1012 
1.0    X  1012        " 

Ethyl  alcohol  
Methyl  alcohol  .  .  . 
Water 

25.0 
31.0 
81   07 

0.3    X  106  (18°  C) 
0.14  X  106 
0  5    X  106 

Linseed  oil  

3.3 

0.61  X  1012        " 

At  very  high  voltage  gradients,  some  evidence  of  a  saturation  range  similar  to  that  of 
gases  has  been  obtained.  With  long  applications  of  voltage,  impure  liquids  undergo  a  so- 
called  electric  cleaning.  This  is  due  to  a  very  low  rate  of  ion  production  so  that  all  ions 
are  swept  to  the  electrodes  and  the  conductivity  drops  nearly  to  that  of  a  pure  liquid. 
Solid  phases  also  are  removed  in  some  cases  by  cataphoresis,  but  the  breakdown  strength 
is  usually  affected  to  a  greater  extent  than  the  conductivity. 

Dielectric  Absorption  and  Losses.  Liquids  exhibit  some  of  the  phenomena  of  dielec- 
tric absorption  shown  by  solids,  but  the  rate  of  decrease  of  the  initial  current  with  time  is 
much  faster  and  normally  is  detectable  only  with  an  oscillograph.  The  characteristic 
"bound  charge"  of  solids  also  is  absent:  the  discharge  current  shows  practically  no  evi- 
dence of  absorption,  except  in  highly  viscous  liquids  of  a  mixed  nature. 

The  initial  high  current  is  reduced,  owing  to  the  accumulation  of  space  charges  in  front 
of  the  electrodes.  The  resultant  non-linear  potential  distribution  can  be  measured  with 
probes;  or  porous  cells  can  be  used  to  remove  the  space  charges  from  the  liquid  for  meas- 
urement. Although  the  absorption  results  in  a  higher  a-c  conductivity,  the  effect  is  of 
much  smaller  magnitude  than  that  in  solids,  and  the  a-c  losses  are  much  lower.  The  non- 
existence  of  any  discharge  phenomena  has  been  attributed  to  the  absorption  of  space 
charges  by  part  of  the  electrode  charges. 

The  power  factor  of  most  commercial  non-polar  insulating  liquids  is  low,  ranging  from 
0.0001  to  0.01.  The  power  factor  at  60  cycles  is  influenced  by  the  d-c  conductivity  and 
may  be  expected  to  double  for  each  10  to  20  deg  cent  rise  in  temperature.  At  high  fre- 
quencies, little  change  in  power  factor  with  temperature  is  to  be  expected  with  true  non- 
polar  dielectrics.  Many  mineral  oils  have  some  polar  impurities  which  produce  character- 
istic peaks  in  the  power-factor  curves  at  frequencies  or  temperatures  where  rapid  changes 
in  dielectric  constant  are  occurring. 

For  an  excellent  summary  of  the  effects  to  be  expected  in  polar  substances,  consult 
Dielectric  Properties  of  Organic  Compounds,  by  S.  0.  Morgan  and  W.  A.  Yager,  Industrial 
and  Engineering  Chemistry,  Vol.  32,  1519  (November  1940). 

Dielectric  Strength.  Breakdown  in  liquids,  like  that  in  gases,  is  not  permanent,  nor 
is  the  subsequent  breakdown  voltage  necessarily  reduced.  Discussion  of  breakdown  must 
be  considered  for  two  cases:  pure  liquids  containing  no  dissolved  or  suspended  gas,  solid, 
or  foreign  liquid;  and  impure  liquids. 

Breakdown  in  pure  liquids  probably  occurs  by  an  ionization  process  similar  to  that  in 
gases  and  undoubtedly  is  aided  by  intense  voltage  gradients  built  up  by  space  charges 
near  the  electrodes.  Change  in  pressure  has  practically  no  effect,  but  increase  in  tem- 
perature decreases  the  breakdown  strength,  particularly  when  the  boiling  point  is  ap- 
proached. The  time  involved  in  the  breakdown  process  may  be  as  short  as  10  ~7  sec  with 
sufficient  overvoltage 

Impure  liquids  usually  break  down  at  much  lower  voltages  for  a  variety  of  reasons,  the 
most  important  of  which  is  the  presence  of  moisture  or  gases.  The  electric  field  tends  to 
liberate  dissolved  gases,  and,  since  the  gas  dielectric  strength  is  only  about  one-tenth  that 
of  the  liquid,  the  gas  ionizes  and  starts  the  discharge.  The  harmful  effects  of  moisture 
and  gases  are  greatly  increased  by  fibers  or  other  suspended  solid  particles  which  absorb 
the  impurity.  Fibers,  particles,  or  foreign  liquids  may  form  "bridges"  or  "chains"  if  the 
dielectric  constant  is  higher  than  that  of  the  liquid.  Sometimes  these  bridges  lead  only 
to  preliminary  or  "pilot"  sparks  which  exert  no  effect  on  the  breakdown.  A  very  small 
percentage  of  impurity  usually  produces  a  marked  lowering  of  breakdown  strength,  but 
larger  percentages  have  only  a  slightly  greater  effect.  Particles  of  carbon  formed  by  arcs 
or  sparks  give  but  a  small  decrease,  which  is  proportional  to  the  concentration. 


2-52 


PROPERTIES  OF  MATERIALS 


Although  the  breakdown  voltage  of  pure  liquids  is  linear  with  distance  in  uniform 
fields,  that  of  impure  liquids  is  considerably  influenced  by  gap  geometry.  A  horizontal 
gap  gives  lower  breakdowns  with  impure  oils  than  a  vertical  gap  because  of  the  difference 
in  the  ease  of  gas  elimination.  Small  gaps  are  quite  liable  to  breakdown  by  fiber  bridges; 
long  needle  gaps  are  scarcely  affected  by  most  impurities.  Increased  gap  area  obviously 
results  in  lower  average  breakdown  with  impure  liquids.  m 

The  effect  of  temperature  on  breakdown  of  impure  liquids  depends  on  the  kind  of 
impurity.  Materials  of  low  dielectric  constant  are  more  readily  expelled  from  the  neld 
as  the  viscosity  is  reduced  by  raising  the  temperature,  but  the  conductivity  of  the  liquid 
is  increased.  Moisture  may  be  expelled,  raising  the  breakdown.  Pressure  increases  the 
dielectric  strength  by  preventing  gas  elimination  or  vaporization  of  the  liquid.  ^ 

Since  formation  of  fiber  bridges  and  gas  elimination  take  an  appreciable  time,  the 
strength  of  liquids  to  transient  voltages  is  little  influenced  by  these  impurities.  With 
steady-state  currents,  the  strength  increases  slightly  as  the  frequency  increases  up  to 
1000  cycles,  but  may  be  as  low  as  30  per  cent  of  the  60-cycle  value  at  radio  frequencies, 
probably  on  account  of  the  heating  effect. 

Commercial  Oils.  For  service  in  which  the  liquid  will  be  in  contact  with  air,  the  use 
of  mineral  oil  has  been  practically  universal  because  of  its  stable  nature  and  low  cost. 
Since  the  breakdown  of  all  oils  is  approximately  30  to  40  kv  rms  in  a  standard  0.1-in.  gap 
between  l-in.-diameter  disks,  little  is  to  be  gained  by  substituting  other  oils  except  a 
higher  dielectric  constant,  which  is  obtained  at  the  cost  of  lower  resistivity.  For  hermet- 
ically sealed  applications,  such  as  capacitors,  purified  castor  oil,  with  a  constant  of  4.7,  is 
often  used.  Carefully  purified,  chlorinated  hydrocarbons,  such  as  "Pyranol,"  with^a 
constant  of  4.5,  are  also  employed  for  this  purpose  and  have  an  effective  advantage  in 
being  explosion-proof. 

Mineral  oils  must  be  carefully  purified  to  remove  unsaturated  compounds  which  cause 
accelerated  oxidation  in  service,  resulting  in  low  resistivity,  low  dielectric  strength,  high 
power  factor,  and  the  rapid  development  of  sludge.  Too  drastic  a  purification,  however, 
removes  naturally  occurring  antioxidants  in  the  oil,  and  stability  is  decreased.  The  oxida- 
tion of  transformer  oils  may  be  avoided  by  the  use  of  oxygen-free  atmospheres  above  the 
oils  as  in  the  "Inertaire"  system. 

Filtration  through  diatomaceous  earth  is  effective  in  increasing  the  resistivity  of  many 
liquids.  Filtration  through  hard  papers  is  commonly  used  for  purifying  and  drying  oils  in 
transformer  service.  Oil  should  be  filtered  when  the  dielectric  strength  in  the  standard 
0.1-in.  gap  drops  below  22  kv  rms.  A  good  oil  will  show  30  to  50  kv  rms.  Low-viscosity 
oils  seem  to  have  the  highest  dielectric  strength,  although  the  flash  and  fire  points  usually 
are  lower. 

Testing  methods  for  electrical  insulating  oils  have  been  standardized  by  the  American 
Society  for  Testing  Materials;  see  ASTM  D117.  Typical  properties  for  commercial  oils 
are  shown  in  Table  4. 

Table  4.    Properties  of  Commercial  Oils 


Property 

Mineral 
Trans- 
former Oil 

Mineral 
Capacitor 
Oil 

Castor 
Oil 

Density,  average  

0.87 

0.91 

0.96 

Viscosity  at  37.8°  C  (100°  F),  in  Saybolt  seconds,  average  
Flash,  point/,  in  d^g  ofint,  minimum            

57 
133 

100 

149 

1400 

Fire  point-,  in  d^g  nfint,  minim  trm  -...-.        -.,,...,  

148 

170 

-40 

-40 

-15 

Neutralization  number,  in  rag  KOH  per  gram,  maximum  

0.03 
2.2 

0.03 
2  2 

2.0 
4  7 

Power  factor,  at  100°  C  and  1000  cycles 

0  0025  max 

0  01 

Resistivity,  ohm-cm,  at  100°  C  

4  X  1011 

>5  X  1012 

6  6  X  10l° 

Dielectric  strength,  at  25°  C,  in  kv,  minimum  

30 

30 

28 

Coefficient  of  expansion  per  deg  cent  

6.3  X  10~4 

6  3  X  10~4 

Thermal  conductivity,  in  cal  per  sec  per  cm  per  deg  cent  

3  X  10~4 

4.3  X  10~4 

Synthetic  Liquids.  Synthetic  insulating  liquids  of  the  non-inflammable  type  are 
known  as  askarels,  and  a  draft  of  proposed  testing  methods  has  been  published  in  the 
Proceedings  of  the  American  Society  for  Testing  Materials,  Vol.  43,  353.  These  liquids 
consist  of  mixtures  of  various  chlorinated  diphenyls  and  tri-  or  dichlorobenzene  so  ad- 
justed that  the  pour  point  is  reduced  below  service  temperatures.  The  dielectric  constant 
is  about  4.2.  Viscosity  at  100  deg  fahr  is  about  the  same  as  that  of  mineral  transformer 
oils.  Dielectric  strength  is  slightly  higher,  and  the  fact  that  the  liquids  are  non-inn  am- 


GASES  AS  DIELECTRICS 


2-53 


mable  permits  the  use  of  large  transformers  without  fireproof  vaults.  On  account  of  the 
non-explosive  nature  of  the  liquid,  the  air  space  above  it  may  be  sealed  from  the  atmos- 
phere with  a  safety  diaphragm  designed  to  relieve  pressure  if  a  fault  occurs.  Improved 
stability  to  oxidation  and  sludging  is  another  advantage  of  these  liquids.  Trade  names: 
Pyranol,  Inerteen. 

Silicones.  Silicons  fluids  are  a  recent  development  and  are  expensive  (as  of  1949), 
but  they  appear  to  be  most  promising  as  an  insulating  medium.  Silicone  liquids  consist 
of  chains  of  alternate  silicon  and  oxygen  atoms  with  various  organic  groups  attached  in 
pairs  to  the  silicon  atoms.  Those  with  two  methyl  groups  attached  are  dimethyl  silicones 
(Dow-Corning  DC  200  fluids).  The  viscosity  increases  with  the  chain  length.  These 
fluids  are  suitable  for  use  from  —40  deg  fahr  to  400  deg  fahr.  Another  series  (DC  500 
fluids)  is  serviceable  from  —  70  deg  fahr  to  200  deg  fahr.  The  general  advantages  of  sili- 
cone  fluids  are: 

1.  Low  temperature- viscosity  slopes. 

2.  High  flash  and  fire  points. 

3.  Low  volatility  and  negligible  vapor  pressure. 

4.  High  resistance  to  oxidation  and  heat. 

5.  Lack  of  color,  odor,  or  toxicity. 

6.  Low  power  factor  over  a  wide  frequency  range. 

7.  Non-corrosive  to  metals  and  non-solvent  for  rubber  and  plastics. 

The  characteristics  of  these  fluids  are  shown  in  Table  5. 

Table  5.    Properties  of  Some  Liquid  Silicones  (Dow  Corning) 


Fluid 
Type 

Viscosity 
Grade, 
centi- 
stokes  at 

Viscosity 
Temperature 
Coefficient 
/  .       ^210°  F  \ 

Freezing 
Point, 
deg  cent 

Boiling 
Point, 
deg  cent 

Flash 
Point, 
deg 
cent 

Specific 
Gravity 
25°  C/25°  C 

Coefficient 
of  Thermal 
Expansion 

Refrac- 
tive 
Index 

at 

25°  C 

V         FiOO°  F  / 

min 

Unit  =  10  3/°C 

25°  C 

DC  500 

1.0 

0.37 

-86 

j  52760  mm 

37.8 

0.818 

1.451 

.3822 

3.0 

.51 

-70 

ca.  800.5  mm 

107 

.896 

1.170 

.394 

10.0 

.57 

-67 

>20QQ-5  mm 

176 

.940 

1.035 

.399 

50.0 

.59 

-55 

>  2500-5  mm 

282 

.955 

1.00 

.402 

DC  200 

100 

.60 

See  Note  1 

See  Note  2 

315 

.968 

0.969 

.4030 

350 

.62 

« 

« 

329 

.972 

0.966 

.4032 

1000 

.62 

" 

" 

337 

.973 

0.963 

.4035 

Note  1.    Recommended  for  use  above  —40  deg  cent. 

Note  2.    Less  than  2  per  cent  volatile  during  48  hours  at  200  deg  cent. 


Electrical  Properties  of  DC  200  Fluids  at  25  Deg  Cent  and  50  Per  Cent 
Relative  Humidity 


Frequency, 
cycles  per  sec 

Dielectric 
Constant 

Power 
Factor 

103 
106 
108 

2.85 
2.83 
2.81 

0.001 
.002 
.006 

Dielectric  strength,  250-300  volts  per  mil 
Volume  resistivity,  1  X  10U  ohm-cm 


7.  GASES  AS  DIELECTRICS 

Dielectric  Constant.  The  dielectric  constants  of  gases  are  close  to  unity  and  nearly 
independent  of  frequency.  The  change  in  dielectric  constant  of  dry  non-polar  gases  with 
temperature  or  pressure  is  slight  and  may  be  calculated  approximately  from  the  equation 


K-  1  =  A  X 


P 


273  -f  t 
where  A  is  a  constant  (2.12  X  10~4  for  air). 

p  is  the  pressure  in  millimeters  of  mercury. 
t  is  the  temperature  in  degrees  centigrade. 


2-54 


PKOPERTIES  OF  MATERIALS 


Table  6.    Dielectric  Constant  of  Gases 


Gas 

Temperature, 
deg  cent 

Pressure, 
atmospheres 

Dielectric; 

Constant 

Observer 


Air  

0 

1 

.000590 

Boltzmann  1875 

19 

20 

.0108 

Tangl  1907 

u 

40 

.0218 

u             tt 

u 

60 

.0330 

"          " 

u 

80 

.0439 

It                       (C 

a 

100 

.0548 

a              a 

23 

.000530 

Braunmuhl  1927 

G&.rbon  dioxide 

0 

1 

.000985 

Klemencic 

15 

10 

.008 

Linde  1895 

u                     u 

20 

.020 

a              u 

u                    u 

40 

.060 

"              " 

Carbon  monoxide  
Ethylene 

0 
0 

.000690 
.0031 

Boltzmann 

Hydrogen        

0 

.000264 

" 

Methane   

0 

.000944 

« 

0 

.00061 

Oxygen  

0 

.00055 

Conductivity  and  lonization.  The  conductivity  of  gases  at  low  potential  gradients  is 
negligible  in  the  absence  of  ionizing  radiation,  such  as  ultraviolet  light  or  X-rays.  If  a 
sufficient  voltage  gradient  exists,  all  tbe  ions  are  drawn  to  the  electrodes  as  fast  as  they  are 
produced  and  the  very  small  ion  current  is  constant  over  a  considerable  range  of  gradient. 
Increasing  the  gradient  beyond  this  saturation  range  accelerates  the  negative  ions  (or 
electrons)  to  a  velocity  which  is  sufficient  to  expel  electrons  from  neutral  gas  molecules  at 
each  collision. 

If  the  number  of  ions  liberated  by  the  collisions  exceeds  the  number  of  negative  ions 
which  are  lost  by  recombination  to  form  neutral  molecules  and  by  diffusion  out  of  the  field, 
the  collision  process  is  cumulative  and  the  current  increases  continuously  to  breakdown. 
If  the  electric  field  is  uniform,  sparkover  will  occur,  but  if  the  high  voltage  gradients  are 
confined  to  a  small  region,  such  as  the  vicinity  of  a  pointed  electrode,  a  local  discharge, 
known  as  corona,  occurs.  Local  discharges  produce  visible  and  ultraviolet  light  which  is 
effective  in  increasing  ionization  throughout  the  field.  This  internal  photo-ionization 
causes  extremely  rapid  breakdowns  of  air  gaps  at  sufficiently  large  voltage  gradients. 

Corona.  The  production  of  a  corona  discharge  requires  a  considerable  current  which  is 
carried  by  ions  of  lower  velocity  in  the  dark  regions  of  the  field.  A  significant  power  loss 
may  occur  if  the  corona  becomes  appreciable.  Corona  is  objectionable  because  it  causes 
radio-frequency  interference  and  produces  ozone  if  oxygen  is  present.  Ozone  and  the 
ultraviolet  light  from  the  discharge  cause  rapid  deterioration  of  many  solid  dielectric 
materials,  especially  of  rubber.  Corona  is  normally  prevented  by  operation  at  reduced 
voltages  or  by  the  use  of  suitable  corona  shields  which  are  designed  to  produce  a  more 
nearly  uniform  voltage  gradient  throughout  the  air  space. 

Dielectric  Strength  and  Sparkover.  Dielectric  strength  is  the  maximum  potential 
gradient  at  the  instant  of  sparkover  or  at  the  onset  of  corona.  The  gradient  is  determined 
by  the  geometry  and  spacing  of  the  electrodes.  Dielectric  strength  is  influenced  by  the 
nature  and  purity  of  the  gas,  by  the  density  of  the  gas,  and  to  a  lesser  degree  by  the  elec- 
trode material.  When  the  mean  free  ion  path  between  collisions  is  lengthened  by  lowering 
the  gas  density,  the  critical  terminal  velocity  necessary  for  ionization  by  collision  is  at- 
tained with  a  lower  potential  gradient.  The  reduction  in  dielectric  strength  with  decreas- 
ing density  continues  until  the  number  of  atoms  between  the  electrodes  is  so  small  that 
very  few  collisions  occur.  If  the  density  is  still  further  reduced  and  the  electric  field  is 

Ta-Ki    TT     TUT-  •  e      i  •      Tk  *     xi  i  non-uniform,  the  sparkover  will  occur 

Table  7.    ICuumnm  Sparking  PotentuHs          over  some  path  ^  than  ^  short. 

est  distance  between  electrodes.  If 
the  field  is  uniform,  the  sparkover 
voltage  will  increase  as  the  density 
decreases  to  very  low  values,  until,  in 
a  high  vacuum,  gradients  as  high  as 
6000  kv  per  cm  may  be  obtained.  The 

.  minimum  in  the  sparkover  voltage  is 

independent  of  electrode  spacing  for  uniform  fields  and  depends  solely  on  the  nature  and 
purity  of  the  gas  and  on  the  electrode  material.  A  typical  set  of  values  for  various  gases  is 
shown  in  Table  7. 


Gas 

Volts 
(d-o) 

Gas 

Volts 
(d-c) 

Air  

341 

Hydro  ge  n 

278 

Carbon  dioxide  .  . 
Helium  

419 
261 

Nitrogen  
Oxygen  

251 
455 

GASES  AS  DIELECTRICS 


2-55 


Sparkover  voltage  in  uniform  fields  is  a  nearly  linear  function  of  the  product  of  gas 
density  and  electrode  spacing  (Paschen's  law). 

An  empirical  equation  which  reproduces  the  entire  sparkover  curve  in  air  for  uniform 
fields,  including  the  region  around  the  minimum  sparking  potential,  is 


V  = 


293 

273  +t 


3.000  +  loge 


+  300  volts  (crest) 


where  p  is  the  pressure  in  millimeters  of  mercury, 
S  is  the  electrode  spacing  in  centimeters. 
t  is  the  temperature  in  degrees  centigrade. 

The  calculation  of  the  sparkover  voltage  for  non-uniform  fields  is  not  simple;  see  P.  W. 
Peek,  Dielectric  Phenomena  in  High  Voltage  Engineering,  McGraw-Hill  (1929)  for  extensive 
formulas  and  tables  for  this  purpose.  At  spacings  of  the  order  of  one-half  the  sphere  radius, 
the  sparkover  voltage  of  a  gap  between  equal  spheres  is  slightly  higher  than  for  a  uniform 
field,  but  as  the  ratio  of  spacing  to  radius  increases  the  sparkover  voltage  becomes  much 
less  than  for  a  uniform  field.  At  ratios  above  2,  corona  occurs  before  sparkover,  but 
corona  is  not  easily  detected  at  60  cycles  until  the  ratio  is  about  8.  Representative  spark- 
over  voltages  for  sphere  gaps  are  shown  in  Table  8. 

Table  8.    Sparkover  Voltage  in  Rms  Kilovolts 

Barometer  76  cm,  temperature  25  deg  cent 
Diameter  of  Spheres 


Gap  Spacing 

2  cm 

6.25  cm 

12.5cm 

25  cm 

cm 

in. 

NG* 

Gf 

NG 

G 

NG 

G 

NG 

G 

0.2 
0.25 
0.3 
0.4 
0.5 
0.6 
0.7 
0.8 
0.9 
1.0 
1.2 
1.4 
1.5 
1.6 
1.8 
2.0 
2.5 
3.0 
4.0 
5.0 
6.0 
7.0 
8.0 
9.0 
10.0 
12.0 
12.5 
15 
17.5 
20 
22.5 
25 
30 
40 

0.079 
,098 
.118 
.158 
.197 
.236 
.276 
.315 
.354 
.394 
.472 
.551 
.591 
.630 
.709 
.787 
.984 
1.181 
1.575 
1,969 
2.362 
2.756 
3.15 
3.543 
3.937 
4.724 
4.921 
5.906 
6.89 
7.874 
8.858 
9.843 
11.811 
15.748 

5.6 

5.6 

6.5 

6.5 

8.0 
10.3 
12.5 
14.8 
17.0 
18.9 
20.8 
22.6 
25.9 
28.9 

8.0 
10.3 
12.5 
14.6 
16.7 
18,6 
20.2 
21.7 
24.4 
26.4 

12.0 

12.0 

12.0 

12.0 

11 

11 

22.5 

22.5 

22.0 

22.0 

22 

22 

31.5 

31.5 

31.5 

31.5 

32 

32 

28.2 

30  0 

41.0 

41.0 

41.0 

41.0 

42 
52 
61 
78 
96 
112 

42 
52 
61 
78 
94 
110 

57.5 
70.5 
81.0 
89.0 
96.0 
102,0 
107.0 
110.0 

56.0 
66.0 
73.0 
79.0 
83.0 
88.0 
90.5 
93.0 

59.0 
76.0 
91.0 
105,0 
118.0 
130.0 
141.0 
151.0 
167.0 

59.0 
75.0 
89,0 
102.0 
112.0 
120.0 
128.0 
135.0 
147.0 

171 

166 

203 
230 
255 
278 
297 
314 
339 
385 

196 
220 
238 
254 
268 
280 
300 
325 

188.0 
201,0 
213.0 

160.0 
168.0 
174.0 

Values  from  F.  W.  Peek,  Dielectric  Phenomena  in  High  Voltage  Engineering,  3d  ed.,  McGraw-Hill 
Book  Co.  (1929). 

*  NG  =  electrodes  balanced  to  ground, 
t  G      •=  one  electrode  grounded. 


2-56 


PROPEKTIES  OF  MATERIALS 


Table  9  shows  the  influence  of  altitude  on  sparkover  voltage  at  constant  temperature 
for  uniform  fields  in  gaps  of  various  lengths.  The  table  also  is  approximately  correct  for 
closely  spaced  sphere  gaps.  Correction  for  the  lower  air  temperatures  shown  is  seldom 
warranted  unless  the  actual  temperature  of  the  air  in  the  gap  is  known. 

Table  9.    Sparkover  Voltages  at  High  Altitudes 

Ratio  of  sparkover  voltage  to  sea-level  sparkover  in  uniform  fields 


Sparkover  Voltage  at  Constant 

Altitude, 

Standard 

Pressure 

Temperature 

Unit  = 
1000  ft 

Air  Temp, 
0°C 

Relative 
Pressure 

Gap  Spacing,  cm 

mm  Hg 

in.  Hg 

0.01 

0.1 

1.0 

10.0 

0 

+  15.0 

760.0 

29.92 

1.000 

1.000 

1.000 

1.000 

1.000 

5 

+5.1 

632.2 

24.89 

0.832 

0.910 

0.862 

0.85 

0.845 

10 

-4.8 

522.6 

20.58 

0.687 

0.828 

0.742 

0.719 

0.712 

15 

-14.7 

428.8 

16.88 

0.565 

0.759 

0.637 

0.603 

0.593 

20 

-24.6 

349.1 

13.75 

0.459 

0.695 

0.545 

0.505 

0.493 

25 

-34.5 

281.9 

11.10 

0.371 

0.640 

0.468 

0.418 

0.390 

30 

-44.4 

225.6 

8.88 

0.296 

0.592 

0.399 

0.345 

0.331 

35 

-54.3 

178.7 

7.04 

0.247 

0.552 

0.340 

0.283 

0.268 

40 

-55.0 

140.7 

5.54 

0.185 

0.517 

0.291 

0.232 

0.217 

45 

-55.0 

110.8 

4.36 

0.146 

0.488 

0.251 

0.190 

0.  175 

50 

-55.0 

87.3 

3.44 

0.115 

0.465 

0.219 

0.159 

0.  141 

The  dielectric  strength  of  gases  is  subject  to  large  changes  with  impurities.  At  low 
pressures,  0.1  per  cent  of  argon  in  neon  gas  reduces  the  dielectric  strength  by  75  per  cent. 
Mercury  vapor  likewise  lowers  the  sparkover  voltage.  At  atmospheric  pressure,  the 
addition  of  small  amounts  of  carbon  tetrachloride  or  chloroform  vapor  increases  the  di- 
electric strength  by  50  per  cent.  Table  10  shows  the  approximate  relative  dielectric 
strength  of  gases.  Although  Freon  (dichlorodifluoromethane)  has  a  higher  relative  dielec- 
tric strength,  its  vapor  pressure  varies  from  9.3  psi  at  —40  deg  cent  to  139  at  +40  deg 
cent,  whereas  the  pressure  with  sealed-in  nitrogen  increases  by  only  35  per  cent  in  this 
interval.  Furthermore,  if  the  Freon  does  break  down,  its  decomposition  products  are 
corrosive  and  carbon  is  deposited.  Dry,  oil-pumped  nitrogen  is  undoubtedly  the  best 
gas  dielectric  unless  cooling  is  involved,  which  may  necessitate  the  use  of  hydrogen. 

Table  10.    Approximate  Relative  Dielectric  Strength  of  Gases 


Gas 


Pressure  in  Absolute  Atmospheres 


V3 

1 

2 

4 

8 

Nitrogen  

1   0 

1  0 

1   0 

1   0 

1   0 

Air  

0  9 

0  9 

0  9 

0  9 

I   0 

Oxygen  

0.9 

0  8 

0  g 

0  8 

Carbon  dioxide  

0  9 

0  9 

0  9 

0  9 

0  8 

Hydrogen  ,  

0  6 

0  6 

0  6 

0  6 

Freon  

2  4 

2  2 

2  2 

2  2 

Nitrogen  plus  carbon  tetrachloride  vapor  

1   6 

1   5 

1   3 

1   2 

Helium  

0.3 

At  frequencies  above  10  kc  the  dielectric  strength  of  air  decreases  slightly.  Owing  to 
the  low  velocity  of  positive  ions,  they  are  not  swept  to  the  electrodes  during  one  half  cycle, 
so  that  they  remain  to  distort  the  field  on  the  next  half  cycle.  At  frequencies  above  60  kc, 
a  reduction  of  7  to  13  per  cent  in  breakdown  voltage  is  to  be  expected,  but  no  further  de- 
crease occurs  up  to  at  least  several  megacycles.  If  the  gap  is  illuminated  with  ultraviolet 
light,  a  decrease  of  17  to  20  per  cent  is  obtained. 


MAGNETIC  MATERIALS 

By  R.  M.TBozorth  and  R.  A.  Chegwidden 

Of  all  the  common  elements  only  iron,  cobalt,  and  nickel  have  magnetic  properties 
greatly  different  from  those  of  air  or  vacuum.  The  magnetic  materials  in  common  use 
consequently  contain  at  least  one  of  these  elements  and  sometimes  all  three  These 


MAGNETIC  CHAEACTEEISTICS 


2-57 


materials,  called  ferromagnetic,  make  possible  the  operation  of  motors,  generators,  trans- 
formers, and  practically  all  electromagnetic  devices. 

Non-ferromagnetic  materials  are  of  little  importance  on  account  of  their  magnetic 
properties.  They  fall  into  two  classes:  paramagnetic  materials  which  are  but  slightly 
more  magnetic  than  a  vacuum,  and  are  therefore  attracted  weakly  by  the  poles  of  an 
electromagnet,  and  diamagnetic  materials  which  are  repelled  weakly  by  an  electromagnet 
Because  the  magnetic  materials  used  in  communication  are  almost  without  exception 
ferromagnetic  materials,  this  article  will  be  devoted  to  a  description  of  this  class.  They 
may  be  divided  into  "high  permeability"  (or  magnetically  "soft")  and  "permanent 
magnet"  (or  magnetically  "hard")  materials.  Ferromagnetic  materials  are  commonly 
referred  to  as  simply  "magnetic"  materials. 

The  magnetic  properties  of  materials  depend  primarily  on  the  nature  of  the  atoms  which 
compose  them.  In  magnetic  materials  the  atoms  are  small  permanent  magnets  that  owe 
their  magnetic  moments  to  uncompensated  spinning  electrons  lying  in  electron  shells 
inside  the  atom.  For  a  material  to  be  ferromagnetic  these  shells  must  be  incomplete  (i.e., 
have  spaces  for  more  electrons  than  are  present)  and  have  an  excess  of  electrons  spinning 
in  one  direction,  and  the  atoms  must  be  arranged  in  regular  fashion  on  a  space  lattice, 
with  atom  centers  not  too  close  together.  The  important  elements  whose  atoms  fulfil 
these  conditions  occur  in  one  part  of  the  periodic  table — they  are  iron,  cobalt,  and  nickel. 
Gadolinium  has  been  found  to  be  ferromagnetic,  and  manganese  and  chromium  can  also 
give  rise  to  ferromagnetism  when  alloyed  or  chemically  combined  with  the  right  non- 
ferromagnetic  elements.  The  Heusler  alloys  are  composed  of  manganese,  aluminum,  and 
copper;  in  comparison  with  many  of  the  alloys  of  iron,  cobalt,  and  nickel,  they  are  quite 
inferior  from  a  practical  magnetic  standpoint  and  have  had  no  commercial  use. 

Magnetic  properties  depend  also  on  crystal  structure,  state  of  strain,  temperature,  and 
other  factors. 

8.  MAGNETIC  CHARACTERISTICS 

MAGNETIZATION  AND  PERMEABILITY  CURVES.  The  properties  of  magnetic 
materials  are  usually  described  first  of  all  by  a  magnetization  curve  such  as  that  shown  in 
Fig.  1.  Here  the  magnetic  induction,  B,  in  a  ring  sample  is  plotted  against  the  magnetizing 

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He 


IRON 


-3-2-1          0         1          2         345 
MAGNETIZING  FORCE, H,  IN  OERSTEDS 


PERMEABILITY,  fd. 
:>  —  w  c»>  ^  m  o>  -j  o> 

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INDUCTION,  B, IN   KILOGAUSSES,  OR 
MAGNETIZING  FORCE,H,IN    OERSTEDS 


FIG.    1. 


Magnetization   Curve   and  Hysteresis 
Loop  for  Annealed  Iron  Ring 


FIG.  2. 


Typical  Permeability  Curves  for  Hot- 
rolled  4  Per  Cent  Silicon  Iron 


force,  H.  B  is  a  measure  of  the  amount  of  the  magnetization;  it  is  denned  specifically 
below  under  "Definitions."  H  represents  the  magnetizing  force  required  to  produce  the 
magnetic  induction,  B;  it  is  usually  measured  hi  oersteds  or  in  ampere-turns  per  inch.  The 
magnetic  induction  is  sometimes  described  in  terms  of  the  intensity  of  magnetization,  I, 
equal  to  (B  —  H)/&ir.  The  increase  in  induction  due  to  the  material  alone  is  B  —  H, 
sometimes  called  the  intrinsic  induction;  this  quantity  becomes  important  when  the  mag- 
netizing force  is  high,  e.g.,  when  determining  Bs,  the  saturation  induction,  highest  attain- 
able value  of  B  —  H  in  a  material. 

The  ease  with  which  a  magnetic  material  can  be  magnetized  is  measured  by  the  ratio 
B/H,  called  the  permeability,  \i.  Typical  permeability  curves  plotted  against  B  and  H 
are  shown  for  a  sample  of  4  per  cent  silicon-iron  in  Fig.  2. 


2-58 


PROPERTIES  OF  MATERIALS 


In  communication  work,  the  permeability  in  low  fields  is  especially  important.  Figure  3 
shows  the  characteristic  ju,  H  curves  for  several  common  materials ;  in  the  lowest  fields  the 
curves  usually  become  straight  lines. 


«XIO3 


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X0.014-IN.    4-79MO 
PERMALLOY 

^- 

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0.001-  IN.      4  -1 
PERMALLOY 

1  45     | 
PERMALLOY 

9  MO 

°c 

>                   O.Ol                O.O2               0,03              O.C 
MAGNETI 

FIG.  3.    Characteristic  ft  vs  H  Ci 

1 
4.0 

3.5 
3.0 
2.5 
2.0 
1.5 
1.0 

as 

<103 

1 

/ 

1  PERMALLOY 

/ 

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3  PER  CENT 
SILICON  -IRON/ 

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FORCE , H 
0s  JT  Curves  at  Low  Magnetizing  Forces 

Hysteresis  Loops.  Other  important  properties  of  magnetic  materials  are  shown  by 
the  hysteresis  loop  produced  when  B  is  plotted  against  H,  as  H  is  increased  to  a  maximum, 
decreased  to  zero,  increased  to  a  maximum  in  the  negative  direction,  again  reduced  to  zero^ 
and  finally  increased  to  the  first  Tn.fl.yJTmi.Tn  as  shown  by  the  dotted  line  in  Fig.  1.  The 
values  of  B  and  of  H  at  which  this  curve  crosses  the  axes  are  called  respectively  the  residual 
induction,  Br,  and  coercive  force,  Hc,  as  indicated  in  the  figure.  The  area  enclosed  within 
the  hysteresis  loop  is  a  measure  of  the  magnetic  energy  transferred  into  heat  during  the 
cycle  and  is  designated  Wh>  Quantitatively 


in  ergs  per  cm3  when  B  and  H  are  in 

8 


oersteds,  respectively. 


0.12       0.16      0.20 


Fia.  4. 


-0.12    -0.08    -0.04       0          0.04      '0.08 
MAGNETIZING  FORCE,  H 

Minor  Hysteresis  Loops  Shown  at  Various  Points  on  the  Major  Loop  for  a  Specimen  of  4-79 
-Molybdenum  Permalloy 


MAGNETIC  CHARACTERISTICS 


2-59 


Demagnetization  Curve.  That  part  of  the  hysteresis  loop  that  lies  in  the  second 
quadrant,  extending  from  Br  to  Hc  in  Fig.  1,  is  called  the  demagnetization  curve  and  is 
especially  important  for  the  description  of  permanent-magnet  materials.  This  is  de- 
scribed more  fully  in  Article  10  and  Fig.  10.  Hysteresis  loops  which  do  not  have  equal 
excursions  of  H  (and  B}  in  opposite  directions  are  unsymmetric  or  minor  loops.  Some  of 
these  are  shown  in  Fig.  4.  In  a  minor  loop  the  ratio  of  the  total  change  in  B,  AS,  to  the 
total  change  in  H,  AH,  is  called  the  incremental  permeability  MA  =  AB/AH.  In  the  limit 
as  AB  and  AH  approach  0,  the  incremental  permeability  is  the  reversible  permeability,  /^-. 
This  is  sometimes  referred  to  as  superposed  permeability,  and  is  discussed  in  more  detail 
in  Article  15  (Fig.  24). 

Air  Gaps.  When  an  air  gap  is  cut  in  a  closed  magnetic  circuit,  such  as  a  ring  sample, 
magnetic  poles  are  produced  on  either  side  of  the  break,  and  the  apparent  permeability  of 
the  material  is  reduced  because  of  the  reluctance  of  the  gap.  The  effect  of  even  a  small  air 
gap  in  a  high-permeability  circuit  may  be  very  appreciable.  See  Fig.  5,  and  Fig.  19  of 

Article  15. 

xio3 


0      2      4       6      8       10      12     14     16     18     20    22    24   26   28    30    32 
APPLIED    MAGNETIZING    FORCE,  H 

FIG.  5.    Magnetization  Curves  and  Parts  of  Hysteresis  Loops  Showing  Effect  of  0.005-in.  Air  Gap  Cut 
in  a  Silicon  Iron  Ring  2-in.  I.D.  by  3-in.  O.D. 

Eddy-current  Loss.  When  magnetic  fields  are  varied  with  some  rapidity,  as  they  are 
in  most  machinery,  the  material  is  subject  not  only  to  the  hysteresis  loss  already  described 
but  also  to  eddy-current  loss.  This  results  from  the  flow  of  electric  currents  within  the 
material,  induced  by  the  changing  flux  in  them.  They  increase  with  increase  in  the  fre- 
quency, conductivity,  and  permeability  of  the  material.  Since  time  is  required  for  these 
currents  to  build  up  and  to  decay,  the  application  of  a  varying  field  is  accompanied  by  a 
delay  in  the  corresponding  magnetic  induction.  The  summation  of  the  hysteresis  and 
eddy-current  losses  is  frequently  called  the  core  loss  or  iron  loss  of  the  material. 

DEFINITIONS.  The  following  definitions  are  taken  from  ASTM  Specification  A127 
which  may  be  referred  to  for  definitions  of  other  terms  relating  to  magnetic  materials. 
This  list  is  in  alphabetical  order,  for  reference  only. 

Coercive  Force.  Hc.  The  magnetizing  force  required  to  bring  the  induction  to  zero  in  a  magnetic 
material  which  is  in  a  symmetrically  cyclically  magnetized  condition.  The  coercivtty  is  that  property 
of  a  material  measured  by  the  maximum  value  of  the  coercive  force. 

Induction,  Intrinsic.  B^.  The  excess  of  the  induction  in  a  magnetic  material  over  the  induction 
in  vacuum,  for  a  given  value  of  the  magnetizing  force.  The  equation  for  intrinsic  induction  is 

Bt  =  B  —  H 

Induction,  Magnetic  (Magnetic  Flux  Density).  2?.  Flux  per  unit  area  through  an  element  of  area 
at  right  angles  to  the  direction  of  the  flux.  The  cgs  unit  of  induction  is  called  the  gauss  (plural  gausses) 
and  is  defined  by  the  equation: 


Under  a-c  conditions  2?ma 


dA 
:  may  be  calculated  as  follows : 

B  X  108 


where  E  is  in  rms  volts ;  ff  is  the  form  factor. 


2-60  PROPERTIES  OF  MATEKIALS 

Induction,  Normal.  5.  The  limiting  induction,  either  positive  or  negative,  in  a  magnetic  material 
which  is  in  a  symmetrically  cyclically  magnetized  condition. 

Induction,  Residual.  Br.  The  magnetic  induction  corresponding  to  zero  magnetizing  torce  in  a 
magnetic  material  which  is  in  a  symmetrically  cyclically  magnetized  condition.  The  retentimty  is  the 
property  of  a  magnetic  material  measured  by  the  maximum  value  of  the  residual  induction. 

Induction,  Saturation.    B8.     The  maximum  intrinsic  induction  possible  m  a  material. 

Magnetic  Flux.  c|>.  A  condition  in  a  medium  produced  by  a  magnetomotive  force,  such  that  when 
altered  in  magnitude  a  voltage  is  induced  in  an  electric  circuit^  linked  with  the  flux.  The  cgs  unit  of 
magnetic  flux  is  called  the  maxwell  and  is  denned  by  the  equation: 

e=  _tf^X10-8 

at 

where  e  =  induced  emf  in  volts,  and  d^/dt  =  time  rate  of  change  of  flux  in  maxwells  per  second. 

Magnetizing  Force.  H.  Magnetomotive  force  per  unit  length.  The  cgs  unit  is  called  the  oersted 
and  is  defined  by  the  equation: 

*-$ 

where  F  is  in  gilberts  and  I  in  centimeters.    For  a  toroid,  or  at  the  center  of  a  long  solenoid,  the  mag- 
netizing force  in  oersteds  may  be  calculated  as  follows: 


— 

where  J  is  in  amperes  and  I  is  in  centimeters. 

Magnetomotive  Force.  jF.  That  which  tends  to  produce  a  magnetic  field.  In  magnetic  testing 
it  is  most  commonly  produced  by  a  current  flowing  through  a  coil  of  wire,  and  its^  magnitude  is  propor- 
tional to  the  current  and  to  the  number  of  turns.  The  cgs  unit  of  magnetomotive  force  is  called  the 
gilbert  and  is  defined  by  the  equation: 

F  =  QArNI 

where  J  is  in  amperes.    Magnetomotive  force  may  also  result  from  a  magnetized  body. 

Permeability,  a-c.  |iac-  A-c  permeability  is  variously  defined,  and  the  values  obtained  for  a  given 
material  depend  on  the  methods  and  conditions  of  measurement.  As  measured  by  the  Standard 
Methods  of  Test  for  Magnetic  Properties  of  Iron  and  Steel  (ASTM  Designation  A34)  ,  it  is  the  ratio 
of  the  maximum  value  of  induction  to  the  maximum  value  of  the  magnetizing  force  for  a  material  in  a 
symmetrically  cyclically  magnetized  condition. 

It  is  sometimes  defined  as  the  ratio  of  the  rms  flux  density  to  the  rms  magnetizing  force.  Some  of 
the  factors  which  affect  a-e  permeability  are  thickness  of  laminations,  frequency,  and  resistivity. 

Permeability,  Incremental.  HA-  The  ratio  of  the  cyclic  change  in  magnetic  induction  to  the  cor- 
responding cyclic  change  in  magnetizing  force  when  the  mean  induction  differs  from  zero. 

Permeability,  Initial,     \LQ.    The  slope  of  the  normal  induction  curve  at  zero  magnetizing  force. 

•Permeability,  Normal.  (A.  The  ratio  of  the  normal  induction  to  the  corresponding  magnetizing 
force.  In  the  cgs  system  the  flux  density  in  a  vacuum  is  numerically  equal  to  the  magnetizing  force, 
and,  consequently,  the  magnetic  permeability  is  numerically  equal  to  the  ratio  of  the  flux  density  to 
the  magnetizing  force.  Thus: 

B 

»  =  H 

Note:  In  a  non-isotropic  medium  the  permeability  is  a  function  of  the  orientation  of  the  medium, 
since,  in  general,  the  magnetizing  force  and  the  magnetic  flux  are  not  parallel. 

Permeability,  Reversible.  HT.  The  incremental  permeability  when  the  cyclic  change  in  induction 
is  vanishingly  small,. 

9.  HIGH-PERMEABILITY  MATERIALS 

Preparation  and  Heat  Treatment.  In  many  applications  such  as  motors,  generators, 
transformers  and  relays,  it  is  desirable  to  have  materials  of  high  permeability.  The  com- 
mon materials  used  are  iron,  silicon-iron  alloys,  and  iron-nickel  alloys  to  which  various 
other  metals  have  been  added.  Commercial  materials  are  usually  melted  in  open-hearth 
or  electric  furnaces,  poured  to  form  ingots,  then  rolled  to  slabs  and  finally  to  sheets  or 
rods  of  the  required  imensions.  After  fabrication  into  the  final  form  in  which  they  are  to 
be  used,  they  must  be  subjected  to  a  heat  treatment  which  is  appropriate  to  the  particular 
alloy,  in  order  to  develop  their  best  magnetic  qualities.  These  heat  treatments  usually 
consist  of  heating  to  some  temperature  lying  between  800  and  1200  deg  cent  and  cooling 
at  a  definite  rate  to  room  temperature.  Figure  6  shows  magnetization  curves  for  ingot 
iron,  (1)  as  hot  and  cold  rolled  to  final  size,  (2)  as  annealed  at  900  deg  cent  for  1  hour,  and 
(3)  after  heat  treatment  at  the  unusually  high  temperature  of  1400  deg  cent  for  6  hours. 
The  anneal  at  900  deg  cent  may  be  referred  to  as  a  strain-relief  anneal;  that  at  the  higher 
temperature  as  a  purifying  anneal  because  during  the  process  some  of  the  impurities  have 
been  removed  from  the  iron. 

^  In  processes  involving  annealing,  account  must  be  taken  of  phase  transformations  and 
"order-disorder"  phenomena  (orderly  arrangements  of  the  atoms)  like  those  found  in  iron 


HIGH-PERMEABILITY  MATERIALS 


2-61 


and  Permalloy,  respectively.  In  some  materials,  such  as  the  nickel-manganese  alloys,  a 
magnetic  material  may  be  made  non-magnetic  by  cooling  rapidly  from  a  high  temperature 
and  thus  preserving  the  disorderly  distribution  of  nickel  and  manganese  atoms  stable  at 
this  temperature.  In  the  iron-nickel  alloys  containing  50  to  80  per  cent  nickel,  marked 
changes  in  properties  are 
produced  by  annealing 
in  the  presence  of  a  mag- 
netic field. 

Magnetic  Elements. 
Of  the  magnetic  ele- 
ments, iron,  cobalt,  and 
nickel,  iron  is  the  only 
one  used  commercially 
to  any  considerable  ex- 
tent in  unalloyed  form. 
It  is  made  in  large  ton- 
nages for  motors,  gen- 
erators, and  relays.  Its 
characteristic  properties 
are  described  in  Fig.  9 
and  Table  1.  Nickel 
finds  a  limited  use  be- 
cause of  its  magneto- 
strictive  properties,  for 
example  in  supersonic  FIG.  6.  Effect  of  Heat  Treatment  on  the  Magnetic  Properties  of  Iron 
underwater  apparatus. 

Iron-silicon  Alloys.  Next  to  unalloyed  iron,  these  alloys  are  used  in  the  greatest 
quantities,  in  power  transformers,  motors,  generators,  relays,  and  receivers.  The  addition 
of  silicon  increases  the  resistivity  and  so  cuts  down  the  power  loss  due  to  eddy  currents, 
and  it  also  has  some  effect  in  increasing  the  permeability  and  decreasing  the  hysteresis 
loss.  Various  commercial  grades  are  available  containing  up  to  about  5  per  cent  silicon. 
Great  advances  have  been  made  in  the  last  few  years  by  fabricating  the  sheet  by  cold  roll- 
ing instead  of  hot  rolling.  As  with  most  magnetic  materials,  the  iron-silicon  crystals  are 
most  easily  magnetized  along  one  particular  crystallographic  direction,  and  the  cold  roll- 
ing and  associated  heat-treating  processes  are  adjusted  to  orient  the  crystals  so  that  as 
many  as  possible  have  their  directions  of  easy  magnetization  aligned  in  the  direction  of 
rolling.  Material  made  in  this  manner  is  called  grain  oriented.  The  grain-orienting  process 


5        6        7        8        9       10 
MAGNETIZING    FORCE,  H 


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INDUCTION  ,B 

FIG.  7.     Comparison  of  the  M  «s  B  Curves  for  Samples  of  Hot-rolled  and  Grain-oriented  Silicon  Iron 

increases  the  permeability  and  reduces  hysteresis  loss.  The  increase  in  permeability,  how- 
ever, is  effective  only  when  the  sheet  is  magnetized  in  the  direction  in  which  it  is  rolled, 
and  in  most  other  directions  the  permeability  is  lowered.  Grain-oriented  silicon-iron  is 
available  in  grades  containing  up  to  slightly  over  3  per  cent  silicon.  Properties  of  the 
grain-oriented  material  containing  3  per  cent  silicon  are  compared  with  the  hot-rolled 
product  in  Fig.  7. 


2-62  PROPERTIES  OF  MATERIALS 

i 

Iron-nickel  Alloys.  These  alloys  are  used  when  particularly  high  quality  is  desired, 
usually  in  transformers  of  various  kinds  and  in  magnetic  shields.  Their  permeabilities 
are  much  greater  than  those  of  other  alloys,  and  they  have  high  resistivity  and  low  energy 
loss.  The  most  important  binary  alloys  contain  78  per  cent  nickel  and  45  to  50  per  cent 
nickel.  The  former,  called  78  Permalloy  or  Permalloy  A,  has  a  high  initial  permeability 
(about  10,000)  and  a  low  coercivity  (0.05)  and  requires  special  heat  treatment  for  the 
development  of  these  properties.  After  the  usual  anneal  at  1000  to  1100  deg  cent  it  must 
be  cooled  rapidly  in  order  to  develop  maximum  quality.  The  iron-nickel  alloys  containing 
45  to  50  per  cent  nickel  are  useful  in  certain  relays,  transformers,  receivers,  and  other 
apparatus.  They  have  reasonably  high  permeabilities  and  incremental  permeabilities, 
and  their  saturation  inductions  and  resistivities  are  higher  than  those  of  the  78  per  cent 
alloy.  The  cooling  rate  from  the  annealing  temperature  is  not  critical  for  these  alloys. 

Another  development  in  this  field  is  the  grain-oriented  50  per  cent  nickel-iron  alloy, 
originated  in  Germany  and  called  Permenorm  5000-Z.  After  special  heat  treatment,  this 
material  exhibits  hysteresis  loops  which  are  practically  rectangular.  The  grain  orienta- 
tion is  accomplished  by  a  99  per  cent  cold  reduction  before  the  final  heat  treatment. 

When  higher  resistivities  are  required,  other  elements  are  added  to  the  iron-nickel  alloys, 
and  it  is  often  found  that  the  resulting  alloy  has  also  higher  initial  and  maximum  perme- 
abilities. One  of  the  most  useful  of  these  is  Molybdenum  Permalloy  containing  4  per  cent 
molybdenum  and  79  per  cent  nickel.  Similar  properties  are  obtained  in  Mumetal  con- 
taining 2  per  cent  chromium,  5  per  cent  copper,  and  75  per  cent  nickel.  Initial  perme- 
abilities of  20,000  to  30,000  and  maximum  permeabilities  of  about  100,000  are  often  found 
in  these  alloys.  The  molybdenum  Permalloy  must  be  heat-treated  under  non-oxidizing 
conditions  and  preferably  cooled  at  a  definite  rate;  the  Mumetal  must  be  heat-treated  in  a 
hydrogen  atmosphere  for  best  results.  An  alloy  containing  5  per  cent  molybdenum  and 
79  per  cent  nickel  is  called  Supermalloy  and  has  an  initial  permeability  of  50,000  to  150,000 
and  a  maximum  permeability  of  about  1,000,000.  These  properties  are  obtained  by  con- 
trolled melting  of  suitable  raw  materials,  heat  treating  in  hydrogen  at  1300  deg  cent,  and 
cooling  at  a  critical  rate. 

Some  interesting  properties  have  been  obtained  in  alloys  containing  nickel,  iron,  and 
cobalt,  called  Perminvars.  A  typical  alloy  contains  45  per  cent  nickel,  25  per  cent  cobalt, 
and  30  per  cent  iron,  and  its  permeability  is  characteristically  independent  of  magnetizing 
force  over  a  relatively  large  range.  To  retain  this  property,  however,  it  must  never  be 
magnetized  above  this  range.  Above  the  range  of  constant  permeability  the  hysteresis 
loops  have  peculiar  constricted  forms  with  very  low  residual  induction  at  intermediate 
field  strengths.  At  present  these  alloys  are  not  being  used  commercially. 

Iron-cobalt  Alloys.  Alloys  of  iron  with  approximately  35  per  cent  to  55  per  cent 
cobalt  are  remarkable  in  that  the  saturation,  B8,  is  higher  than  that  of  either  iron  or  co- 
balt. One  of  these  alloys,  containing  equal  parts  of  iron  and  cobalt,  is  called  Permendur; 
it  is  useful  where  high  flux  densities  are  necessary  as  in  the  pole  tips  of  electromagnets. 
The  binary  alloys  of  iron  and  cobalt  can  be  hot  rolled  and  machined,  but  they  are  very 
difficult  to  cold  roll.  Vanadium  Permendur  contains  2  per  cent  vanadium  and  equal  parts 
of  iron  and  cobalt;  the  addition  of  vanadium  makes  it  possible  to  cold-roll  the  alloy  to  thin 
sheets.  In  this  form  it  has  found  an  important  application  in  telephone  diaphragms. 
Vanadium  Permendur  can  be  machined  and  even  punched  after  cold  rolling,  but  it  be- 
comes somewhat  brittle  when  annealed.  At  high  inductions  the  superposed  permeability 
of  the  iron-cobalt  alloys  is  higher  than  that  of  any  other  material  (see  Fig.  24) . 

Other  Alloys.  Among  the  other  alloys  that  have  high  permeability  may  be  mentioned 
aluminum-iron,  molybdenum-iron,  and  Sendust,  the  last  developed  by  the  Japanese. 
Aluminum-iron  containing  up  to  about  6  per  cent  aluminum  has  been  produced  having 
magnetic  properties  somewhat  better  than  the  iron-silicon  alloys,  but  because  of  manu- 
facturing difficulties  it  has  never  been  popular.  In  Japan  high  permeabilities  have  been 
obtained  in  the  alloy  containing  14  to  15  per  cent  aluminum,  called  Alfer.  Iron-molyb- 
denum alloys  also  have  good  magnetic  permeabilities  at  low  and  moderate  flux  densities 
and  are  more  ductile  than  the  iron-silicon  alloys.  Although  they  are  not  in  common  use, 
at  teast  one  manufacturer  is  planning  to  market  them.  Sendust  contains  about  9  per  cent 
silicon,  5  per  cent  aluminum,  and  the  rest  iron.  Both  initial  and  maximum  permeability 
are  high,  the  initial  sometimes  as  high  as  35,000.  Sendust  is  an  unusual  high-permeability 
material  in  that  it  is  quite  brittle  and  must  be  cast  and  ground  to  finished  size.  Because 
of  its  brittleness,  Sendust  has  been  formed  into  powder  and  used,  especially  in  Japan,  in 
pressed  powder  cores  for  loading  coils  and  other  coils  for  high-frequency  circuits 

Other  materials  are  useful  in  powdered  form  for  applications  of  this  kind.  Most  impor- 
tant among  them  are  powdered  Permalloy  and  Carbonyl  iron.  The  former  has  the  nominal 
composition  of  2  per  cent  molybdenum,  80  per  cent  nickel,  and  18  per  cent  iron;  it  is 
melted  without  the  addition  of  a  deoxidizer  or  desulfurizer,  and  after  hot  rolling  it  can 


HIGH-PERMEABILITY   MATERIALS 


2-63 


2-64 


PROPERTIES  OF  MATERIALS 


be  crushed  to  a  fine  powder.    It  is  then  mixed  with  a  small  amount  of  insulation  and  pressed 
into  a  solid  core  and  heat-treated.    The  permeability  varies,  depending  upon  the  amount 

of  insulation,  from  about  10  to  125.  Both 
hysteresis  and  eddy-current  losses  are  ex- 
tremely small,  and  the  permeability  is  con- 
stant over  a  large  range  of  magnetizing  force. 
Carbonyl  iron  powder,  prepared  by  de- 
composing iron  carbonyl,  Fe(CO)5,  in  the 
vapor  phase,  consists  of  small  spheroids 
which  are  later  insulated  and  pressed.  All 
these  powdered  materials  are  used  in  the 
cores  of  coils  operating  at  high  frequency; 
Fig.  8  shows  some  of  their  properties. 

Often,  when  the  frequency  is  greater  than 
1  megacycle,  a  powder  is  used  composed  of 
one  of  the  iron  oxides  Fe2Os  or  FegO^  Also, 
properly  treated  mixtures  of  complex  f  errites, 
e.g.,  manganese  and  zinc  ferrites,  have  been 
found  to  have  very  high  electrical  resistivities 
and  initial  permeabilities  of  the  order  of  1000. 
Materials  of  this  type,  developed  in  Holland 
under  the  name  Ferroxcube,  have  been  used 
advantageously  for  certain  types  of  high- 


PERMEABILITY,  JUL 

^o8&888§§ 

"\ 

/ 

V  ^ 
\ 

y 

^x 

X 

—  * 

" 

MOLYBDENUM 
PERMALLOY 
POWDER 

CARBONYL  IRON 
TYPE    55 

^ 

** 

0     20    40       100  200             1000  2OOO       10.OC 

INDUCTIONS 

FIG.    8.     Examples    of  Permeability  Curves   for 
Cores  of  Molybdenum  Permalloy  and  Iron  Car- 
bonyl Powders 


frequency  coils.    The  ferrites  in  general  have  much  higher  initial  permeabilities  but  less 
stability  with  temperature  as  compared  with  the  insulated  powder  materials. 

Many  of  the  properties  of  the  most  important  commercial  magnetic  materials  of  high 
permeability  are  collected  in  Table  1  and  Fig.  9. 


PERMEABILITY,  JJL 


o.oot  0002  coo*,     aoi    002   ao4      o.i     02     04  o.e   1.0 


FIG.  9.    Magnetization  and  Permeability  Curves  for  Magnetic  Materials  in  Common  Use 

Non-magnetic  Materials.  It  is  often  desirable  in  dealing  with  magnetic  circuits  to 
have  a  steel  for  structural  purposes  which  is  non-magnetic.  One  commonly  used  material 
of  this  sort,  called  Lomu,  contains  about  10  per  cent  manganese,  8  per  cent  nickel  and 
the  remainder  iron;  chromium  and  silicon  are  also  sometimes  added.  Another  material 
is  of  the  stainless-steel  type  containing  20  to  25  per  cent  nickel  and  25  per  cent  chromium. 


PERMANENT-MAGNET  MATERIALS 


2-65 


10.  PERMANENT-MAGNET  MATERIALS 

Permanent  magnets  are  useful  for  their  ability  to  maintain  a  magnetic  field  in  space 
without  the  aid  of  an  external  source  of  power.  Since  there  is  no  heating,  and  since  modern 
alloys  can  produce  very  high  fields,  they  are  used  extensively  in  loudspeakers,  small  gen- 
erators, etc.  The  properties  of  these  materials  are  best  described  by  means  of  the  de- 
magnetization curve  already  mentioned.  Typical  curves  for  many  of  the  common  mate- 
rials are  shown  in  Fig.  10.  In  evaluating  materials  it  is  also  desirable  to  use  an  energy 


12 
11 
10 
9 

T>8 

CD 

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I6 

5 
4 
3 
2 
1 
n 

,  — 

^-  — 

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f 

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R6M, 

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/ 

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8OO  600  400 

DEMAGNETIZING  FORCE, 


t  2  3 

ENERGY  PRODUCT, 


4X106 


FIG.  10.  Demagnetization  and  Energy  Product  Curves  of  Some  Important  Permanent-magnet  Mate- 
rials, The  materials  corresponding  to  the  abbreviations  may  be  recognized  by  reference  to  Table  2. 

product  curve  formed  by  plotting  the  product  of  B  and  H,  BH,  against  the  magnetic 
induction  B,  for  points  on  the  demagnetization  curve;  this  is  plotted  also  in  Fig.  10.  Such 
a  curve  is  a  measure  of  the  energy  that  can  be  stored  in  the  magnet,  and  the  best  single 
criterion  for  the  value  of  a  permanent-magnet  material  is  the  maximum  value  of  this 
product,  designated  (BH)m-  In  the  design  of  magnetic  circuits  involving  permanent 
magnets  an  attempt  is  usually  made  to  have  the  magnetic  induction  in  the  magnet  equal 
to  the  B  for  which  (BET)  is  equal  to  (BH}m. 

The  magnetic  behavior  of  a  magnet  may  be  described  by  reference  to  Fig.  11.  The 
line  OA  depends  on  the  dimensions  of  the  magnet  and  is  called  the  load  line;  it  is  fixed  by 
the  demagnetizing  action  of  the  air  gap  in  the  magnetic  circuit.  When  the  external  field 
has  been  removed,  B  and  H  will  be  determined  by  some  point  on  this  line,  preferably  the 
point  for  which  (BH)  is  a  maximum.  It  is  common  practice  to  "stabilize1*  a  magnet  by 
applying  a  small  negative  magnetizing  field  (point  C)  and  then  removing  it  (point  D). 
Extraneous  disturbing  fields  will  then  cause  changes  in  induction  corresponding  to  minor 
loops  such  as  CDEC.  The  minor  loops  in  the  third  quadrant,  under  the  demagnetizing 
curve,  have  slopes  approximately  the  same  as  the  slope  of  the  demagnetizing  curve  just 
below  the  point  B;  they  are  important  in  predicting  the  changes  in  induction  that  occur, 
e.g.,  in  generators. 

Permanent-magnet  materials  may  be  classified  under  the  following  headings: 

Carbon  steel  with  or  without  alloying  elements. 

Dispersion-hardened  alloys. 

Types  heat-treated  in  a  magnetic  field. 

Ductile  alloys. 

Powdered  materials. 

Miscellaneous  special  materials 


2-66 


PROPERTIES  OF  MATERIALS 


4 
Hd 

FIG.  11.    Curves  Useful  in  the  Design  of  Permanent  Magnets.    See  text. 


60O  4OO  2OO 

DEMAGNETIZING  FORCE,  H 


23 
ENERGY  PRODUCT, 


5XI06 


Carbon  Steels.  The  carbon  steels  are  so  called  because  they  depend  upon  carbon 
compounds  for  their  permanent-magnet  qualities.  These  materials  are  usually  prepared 
by  hot  rolling  to  finished  size  and  quenching  from  about  800  to  950  deg  cent  in  water  or  oil. 
Their  permanent-magnet  characteristics  tend  to  deteriorate  with  time,  and  it  is  customary 
to  pre-age  such  magnets  before  use  by  maintaining  them  for  many  hours  at  temperatures 
between  100  and  150  deg  cent.  Although  their  magnetic  properties  are  not  of  very  high 
quality,  many  of  the  carbon-steel  magnet  alloys  are  very  useful  because  of  their  low  cost. 

The  most  common  steels  are  carbon-manganese,  chromium,  tungsten,  and  cobalt  steels* 
There  is  a  great  variety  of  materials  of  this  kind  containing  various  percentages  of  the 
alloying  elements  and  about  0.8  per  -cent  carbon.  Representative  alloys  have  been  in- 
cluded in  Table  2.  The  coercive  force  and  energy  product,  as  well  as  the  cost,  usually 
increase  with  the  alloying  element.  The  highest  energy  product  obtained  in  this  group  is 
about  1.0  X  106. 

Dispersion-hardening  Allays.  These  alloys  contain  no  essential  carbon  but  depend 
for  their  hardening  upon  the  precipitation  of  one  solid  phase  in  another.  They  are  ordi- 
narily heated  to  1300  deg  cent  and  quenched  in  air  or  oil  and  are  subsequently  maintained 
at  600  to  700  deg  cent  for  several  hours;  they  are  therefore  often  referred  to  as  "age- 
hardening"  alloys.  Generally  they  are  quite  brittle  and,  with  the  exception  of  one  type,, 
must  be  cast  and  ground  to  final  size.  One  of  the  first  alloys  of  this  type  to  be  used  com- 
mercially contains  71  per  cent  iron,  12  per  cent  cobalt,  and  17  per  cent  molybdenum;  it 
is  called  Remdttoy  or  Comol.  This  alloy  may  be  hot-rolled  and  machined  to  required  size 
like  the  carbon  steels.  To  give  it  permanent-magnet  qualities,  it  is  quenched  from  1200 
deg  cent  and  aged  at  700  deg  cent  for  about  an  hour.  Like  most  cuspersion-hardening 
alloys,  the  properties  of  Remalloy  do  not  change  appreciably  with  time.  It  has  found 
use  in  meters,  receivers,  and  other  devices. 

Large  quantities  of  dispersion-hardening  alloys  are  made  of  the  iron-nickel-aluminum 
type  to  which  have  been  added  cobalt,  copper,  or  titanium.  All  these  alloys,  called  in 
this  country  the  Alnicos,  are  brittle  and  must  be  cast  and  ground  to  size.  The  addition  of 
titanium  up  to  S  per  cent  is  sometimes  effective  in  causing  high  coercive  force.  Heat 
treatments  and  properties  are  given  in  Table  2. 

Types  Heat-treated  in  a  Magnetic  Field.  The  most  important  of  the  dispersion- 
hardening  alloys  was  developed  in  Holland  as  TiconaL  and,  with  slight  variations,  is 
known  in  the  United  States  as  Alnico  5  and  in  England  as  Alcomax.  The  alloy  of  the 
proper  composition  as  given  in  the  table  is  heated  to  1300  deg  cent  and  then  cooled  in  air 
in  the  presence  of  a  strong  magnetic  field  which  must  be  applied  to  the  magnet  in  the 
direction  in  which  the  best  properties  are  desired.  After  aging  at  about  600  deg  cent 
energy  products  as  high  as  5  X  106  are  obtained. 


PERMANENT-MAGNET  MATERIALS 


2-67 


£•§  a 
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2-68 


PROPERTIES  OF  MATERIALS 


The  interesting  oxide  magnets  made  by  the  Japanese  also  require  a  strong  magnetic 
field  for  developing  their  best  properties.  The  proper  amounts  of  iron  and  cobalt  oxides 
(Fe203  Fe304,  and  Co203)  are  pressed  and  heated  to  1000  deg  cent.  They  are  then 
cooled  and  crushed  and  pressed  to  final  form,  then  reheated  to  1000  deg  cent,  and  finally 
cooled  in  a  strong  magnetic  field.  Large  variations  are  found  in  coercivity  and  retentiyity, 
depending  on  how  the  oxides  are  mixed  and  treated;  Hc  ranges  from  600  to  1000  and  #r 
from  4000  to  1500.  This  material  is  unusually  light  in  weight  and  has  exceptionally  high 
resistivity.  One  product,  marketed  under  the  trade  name  Vectohte,  has  a  coercivity  of 
900  and  a  retentivity  of  1600. 

Ductile  Alloys.  Within  the  last  few  years  a  number  of  ductile  permanent-magnet 
alloys  have  appeared  on  the  market.  The  largest  application  has  been  m  wire  and  tape 
form  for  the  magnetic  recording  of  speech.  The  first  alloys  of  this  type  were  based  on  the 
German  material  containing  20  per  cent  iron,  20  per  cent  nickel,  and  60  per  cent  copper, 
known  in  this  country  as  Cunife.  Another  variation,  called  Cunico,  contains  cobalt. 
Most  of  these  alloys  can  be  cold-drawn  to  fine  wires;  in  fact,  such  cold  reduction  is  often 
necessary  to  develop  their  best  properties.  Viccdloy,  made  of  iron,  cobalt,  and  vanadium, 
is  another  alloy  of  this  type.  Its  properties  also  depend  upon  the  amount  of  cold  reduction, 
and  an  energy  product  of  1.5  X  106  can  be  developed  after  cold  rolling  and  annealing  at 
about  700  deg  cent.  By  drastic  cold  reductions  energy  products  as  high  as  4  X  10  can 
be  obtained. 

Powdered  Materials.  Several  of  the  brittle  alloys  mentioned  above  can  be  produced 
from  powders  which  are  pressed  into  the  desired  shape,  sintered  at  a  high  temperature, 
and  heat-treated  to  give  the  best  magnetic  properties.  This  is  often  advantageous  in 
producing  small  magnets,  when  the  methods  of  powder  metallurgy  can  be  used.  Alnico  2 
is  important  in  this  class,  and  Alnico  5  has  been  used  on  an  experimental  scale. 

A  recent  development  is  material  obtained  by  pressing  powder  of  very  fine  size,  such 
as  that  produced  by  the  reduction  of  iron  compounds  at  low  temperature.  One  product, 
manufactured  in  France,  is  composed  essentially  of  fine  particles  of  iron,  sometimes  with 
admixture  of  cobalt,  and  has  properties  much  like  those  of  Alnico  2. 

Special  Alloys.  A  few  other  alloys  are  worthy  of  brief  mention.  These  are  the  cobalt- 
platinum  alloys,  the  manganese-silver-aluminum  alloy  (Silmanal),  and  certain  alloys 

formed  by  electrodeposition  on  a  mercury  sur- 
face, e.g.,  iron  with  a  small  admixture  of  zinc 
or  iron-cobalt-nickel-aluminum  alloys.  Ex- 
perimentally this  alloy  has  been  made  with 
coercivities  of  400  to  500  and  retentivities  of 
9000  to  11,000. 


£4 


SATURATION,  85 


rPLm 


,/Mo 


UPPER 
PORTION 


MIDDLE 
PORTION 


LOWER  PORTION 


11.  MAGNETIZATION  CURVE 


In  this  section  will  be  considered  briefly  the 
nature  of  the  changes  in  magnetization  that 
correspond  to  different  parts  of  the  magneti- 
zation curve,  and  some  relations  valid  in  each 
part. 

The  magnetization  curve  may  be  divided 
into  three  parts  separated  by  the  "instep"  and 
"knee"  (see  Fig.  12).  At  the  instep  the  curve 
takes  a  sudden  upward  turn  and  the  perme- 
ability increases  rapidly.  At  the  knee  this 
trend  is  reversed  and  the  curve  becomes  more 
and  more  horizontal  and  approaches  asympto- 
tically to  saturation.  In  each  of  the  three 
parts  of  the  curve,  magnetization  proceeds  by 
a  different  mechanism,  as  described  below. 

A  magnetic  material  is  composed  of  many 
small  magnetized  regions  or  domains,  each  of 
which  is  always  magnetized  to  saturation  in 
some  one  direction.  When  the  material  as  a 
whole  is  unmagnetized,  these  domains  are  ar- 
ranged in  various  directions  so  that  the  net  magnetization  of  the  material  is  zero.  The 
effect  of  the  field  is  to  change  either  the  direction  in  which  the  domains  are  magnetized  or 
to  change  the  volume  of  some  of  the  domains  at  the  expense  of  their  neighbors.  This  may 
be  made  clear  by  reference  to  Fig.  13,  where  the  domains  are  represented  by  arrows  indi- 


O  0.1  0.2  0.3  0.4  0^ 

MAGNETIZING  FORCE,  H 

FIG.!  12.     Magnetization    Curve   Showing 

the  Points  of  Special  Interest,  and  Division 

into  Three  Main  Parts 


EFFECT  OF  TEMPERATURE 


2-69 


eating  the  directions  in  which  they  are  magnetized.  The  directions  of  stable  magn^etiza- 
tion  are  determined  in  an  annealed  material  by  the  magnetic  properties  of  the  crystals  of 
which  it  is  composed,  and  in  severely  cold- worked  material  by  the  internal  strains  present. 
In  the  first  part  of  the  magnetization  curve,  below  the  instep,  magnetization  proceeds 
by  small  displacements  of  the  boundaries  between  domains,  a  process  illustrated  in  part 
(6)  of  Fig.  13.  In  this  portion  the  permeability  usually  increases  linearly  with  magnetiz- 
ing force: 

M  =  a  -j-  bH 

In  the  second  part  of  the  curve,  between  the  instep  and  the  knee,  the  domains  change 

direction  suddenly  and  the  magnetization  changes 
from  one  direction  of  stable  magnetization  to  an- 
other. In  this  section  Steinmetz*  law  of  hysteresis 
is  applicable: 

Wh  =  T/S1-6 


(b)     PARTIAL  MAGNETIZATION 


(C)  SUDDEN   REVERSALS  COMPLETE 
(KNEE  OF   MAGNETIZATION  CURVE) 


*  V 


(d)  SATURATED,  DOMAINS    ROTATED 
IN   HIGH  FIELD 


CRYSTAL 
AXES 


MAGNETIC 
FIELD 


200  400  600 

TEMPERATUR£,eC 


800 


FIG.  13.    Diagram  Illustrating  Changes 
in  Domain  Structure  with  Magnetiza- 
tion in  a  Single  Crystal  of  Iron 


FIG.  14.    The  Magnetic  Induction  of  Iron  Meas- 
ured at  Various  Temperatures,  with  Various  Im- 
pressed Fields 


In  the  third  section  the  domains  rotate  smoothly  from  the  stable  directions  indicated  in 
(c)  into  parallelism  with  the  magnetic  field  as  indicated  in  (d) ;  here  the  Frohlich-Kennelly 
relation 

i  -  c  +  dH 

M 
is  approximately  valid. 

Evidence  for  the  domain  structure  of  materials  is  found  in  the  Barkhausen  effect,  which 
proves  that  sudden  changes  in  magnetization  occur  in  the  middle  section  of  the  magnetiza- 
tion curve,  and  in  the  existence  of  powder  patterns  which  can  be  seen  under  a  microscope 
and  which  show  that  the  magnetic  field  at  the  surface  of  a  demagnetized  magnetic  mate- 
rial varies  from  place  to  place  over  regions  about  0.1  mm  apart. 


12.  EFFECT  OF  TEMPERATURE 

The  magnetization  of  a  material  can  be  altered  not  only  by  changing  the  magnetic  field 
but  also  by  varying  mechanical  stress  or  temperature.  Temperature  affects  the  magnetic 
properties  of  all  materials,  in  a  way  that  depends  on  the  induction  and  the  character  of 
the  material.  Figure  14  shows  this  effect  for  iron.  Eventually,  as  the  temperature  is 
raised,  the  material  becomes  non-magnetic,  the  temperature  at  which  this  occurs  being 


2-70 


PROPERTIES   OF  MATERIALS 


called  the  Curie  point  of  the  material.  When  a  high  constant  magnetic  field  is  present,  the 
magnetization  decreases  continually  as  the  temperature  increases,  and  at  a  faster  and 
faster  rate  as  the  Curie  point  is  approached.  When  a  low  field  is  present,  the  permeability 
first  increases  with  temperature  and  then  decreases  again  and  approaches  1  at  the  Curie 

For  some  applications  it  is  desirable  to  have  a  material  with  a  permeability  that  de- 
creases rapidly  as  the  temperature  increases.  These  materials  are  used  in  compensating 
permanent  magnets  for  changing  temperature  and  for  stabilizing  pressed  powdered  cores 
to  make  their  inductance  independent  of  temperature.  The  alloys  used  for  this  purpose 
have  a  Curie  point  near  room  temperature  so  that  the  material  loses  its  magnetism  rapidly 
as  the  temperature  increases  in  this  region.  One  such  alloy  contains  about  30  per  cent 
nickel  and  the  rest  iron;  it  has  been  used  for  compensating  permanent  magnets.  Another 
type,  containing  80  per  cent  nickel,  12.5  per  cent  molybdenum,  and  the  rest  iron,  is  com- 
monly used  for  stabilizing  pressed  powdered  Permalloy  cores.  Still  another  type  contains 
about  35  per  cent  nickel,  5  per  cent  chromium,  0.3  per  cent  silicon,  and  the  rest  iron. 


13.  STRESS  AND  MAGNETOSTRICTION 

Figure  15  shows  the  way  in  which  the  magnetization  may  be  affected  by  the  application 
of  tensile  stress.  A  tension  well  within  the  elastic  limit  of  the  material  will  increase  the 
magnetization  of  45  Permalloy  and  decrease  that  of  pure  nickel.  Under  certain  conditions 
the  permeability  of  some  materials  is  increased  by  a  factor  of  50  by  a  stress  within  the 
elastic  limit.  Materials  whose  permeabilities  are  increased  by  tension  are  said  to  have 
positive  magnetostriction,  because  they  expand  a  few  parts  per  million  when  they  are 


TENSION  =2800  LB/1N.2 

r— i r 


234 

MAGNETIZING    FORCE,  H 


-30 


7 
I 


NICKEL 


FIG.  15.    Magnetization  Curves  for  45  Permalloy  and 
Nickel  as  Affected  by  Tension 


40      80      120     160    200   24O    280 
MAGNETIZING   FORCE  ,  H 

FIG.  16.    Fractional  Change  in  Length  with 
Magnetizing  Force  for  45  Permalloy  (Posi- 
tive Magnetostriction)  and  for  Nickel  (Neg- 
ative Magnetostriction) 

magnetized;  conversely,  the  permeability  is  decreased  by  tension  if  magnetization  causes 
contraction  of  the  material  (negative  magnetostriction,  as  in  nickel).  Some  materials, 
like  iron,  have  positive  magnetostriction  in  low  fields  and  negative  magnetostriction  in 
high  fields. 

Figure  16  shows  how  46  Permalloy  and  nickel  change  in  length  as  the  field  strength 
increases.  Such  magnetostriction  is  capable  of  converting  magnetic  energy  into  mechanical 
energy,  and  nickel  is  often  used  in  magnetostriction  oscillators  to  produce  supersonic 
vibrations  in  air  or  under  water,  where  they  are  effective  in  sound  ranging.  The  Japanese 
have  used  a  new  alloy,  Alfer,  containing  13  per  cent  aluminum  and  the  rest  iron,  for  mag- 
netostriction oscillators;  its  magnetostriction  is  about  the  same  in  magnitude  as  that  of 
nickel,  but  is  opposite  in  sign. 


14.  EFFECT  OF  FREQUENCY 


_    When  the  field  acting  on  a  magnetic  material  is  alternated  rapidly,  eddy  currents  are 
induced  in  the  material.  They  act  so  as  to  keep  the  a-c  field  from  penetrating  effectively 


EFFECT  OF  FREQUENCY 


2-71 


more  than  a  certain  distance  below  the  surface;  this  distance  is  measured  in  a  rough  way 
by  the  expression:  __ 


wherein  jj,  —  true  permeability  (d-c)  . 

p  =  103  times  resistivity  in  microhm-centimeters. 
/  =  frequency  in  cycles  per  second. 

The  properties  of  the  material  are  affected  in  three  ways:  (1)  the  effective  permeability 
is  reduced,  (2)  the  energy  loss  in  the  material  is  increased,  and  (3)  there  is  a  time  lag 
between  the  magnetizing  force  and  the  corresponding  induction. 

Only  under  certain  restricted  conditions  can  the  effects  of  alternating  current  be  pre- 
dicted with  any  assurance  of  correctness.  When  the  permeability  is  constant,  as  it  usually 
is  only  when  B  is  very  small,  the  effective  permeability,  £  (as  determined  with  an  induc- 
tance bridge)  is  given  by  the  relation  : 


sinh  6  +  sin  d 


H       6     cosh  6  +  cos  d 
in  which  6  =  Z-rrtVuf/p  —  t/s. 

t  —  thickness  of  sheet  in  centimeters. 

Effective  permeabilities,  calculated  by  means  of  this  equation,  are  compared  with  actual 
measurements  in  Fig.  17. 


20' 

18 

16 
(3. 

>> 

Zi 
512 

< 

UJ 

iio 

s 

ge 

i' 

u. 

UJ 

4 

2 
n 

do2 

1^-. 

-„.. 

^~^< 

^^ 

__ 

^*s« 

—  -«. 

X 

X; 

x; 

^ 

N 

N 

\ 

V 

Y\ 

\  \ 

X 

^ 

\ 

\ 

\ 

\ 

\ 

45  PERMALLOY 
THEORETICAL 
ACTUAL 

\ 

"f\ 

\ 

^ 

\ 

•v^J 

'4 

k 

""v. 

Si  . 

**•*. 

\ 

^sv 

-^ 

^ 

"*s. 

^•^, 

-^^. 

—  «. 

>~» 

0,04  0.1  0.2  0.4     0.6  a8  I  2         34         6      8   10  20      30  40      60  80100 

FREQUENCY  IN    KILOCYCLES    PER   SECOND 

FIG.  17.    Change  in  Effective  Permeability  with  Frequency  for  45  Permalloy  of  Different  Thicknesses 
When  B  ^>  1  (at  high  frequencies)  this  expression  reduces  to: 

1         jos 


Similar  expressions  are  applicable  to  specimens  in  the  form  of  wire  or  cylinders. 

When  /x  is  constant  and  6  <  1  (frequency  and  induction  low)  the  power  loss  may  be 
expressed  by  the  equation: 

-  =  27r/(^o  +  hB  +  ef) 
Ju 

in  which  R  =  excess  of  a-c  resistance  (by  a-c  bridge)  over  d-c  resistance,  in  ohms. 
L  =  inductance  of  coil  in  henrys. 
B  =  maximum  induction  in  gausses. 

This  relation  is  especially  adapted  to  materials  used  in  communication  circuits  in  which 
there  are  feeble  alternating  currents.    The  constants  h  and  e  measure  the  hysteresis  and 


2-72 


PROPERTIES  OF  MATERIALS 


eddy-current  losses,  respectively,  and  ho  is  of  unknown  origin,  important  only  at  the  low- 
est B's. 

At  low  frequencies  (6  <  1)  and  high  inductions  (5  =  1000  to  14,000  in  silicon-iron),  the 
power  loss  in  ergs  per  centimeter3  per  second  is: 

W  =  rjB^f  +  eB2/2 

The  hysteresis  constant,  77,  and  the  eddy-current  constant,  e,  can  be  determined  in  an 
approximate  way  by  plotting  W/f  vs.  /  for  given  values  of  B.  At  high  frequencies  eddy- 
current  loss  is  usually  more  important  than  hysteresis  loss,  and  is  given  in  ergs  per  cen- 
timeter3 per  second  by  the  relation: 


in  which  J-?  is  the  value  of  B  averaged  over  the  cross-section  of  the  sheet. 

At  frequencies  higher  than  108  cps  the  true  permeability,  ju,  of  magnetic  materials 
begins  to  decrease  substantially,  approaching  a  value  of  the  order  of  1  at  frequencies 
around  109  to  1011  cps. 

Magnetization  also  affects  the  resistivity  of  magnetic  materials.  The  change  is  almost 
invariably  an  increase  in  resistivity  with  magnetization,  the  amount  of  the  increase  vary- 
ing from  less  than  1  per  cent  to  about  5  or  10  per  cent  at  room  temperature,  and  even  more 
at  low  temperatures.  Similarly  Young's  modulus  may  be  changed  by  about  10  per  cent 
by  magnetizing  to  saturation.  Unusual  variations  are  also  observed  in  specific  heat, 
thermal  expansion,  and  other  physical  properties  of  magnetic  materials. 


15.  MEASUREMENT  OF  MAGNETIC  CHARACTERISTICS 

Although  it  requires  many  different  measurements  to  determine  all  the  magnetic  char- 
acteristics of  a  material,  the  most  important  properties  can  be  obtained  from  a  magnetiza- 
tion curve  and  hysteresis  loop,  an  a-c  measurement  of  the  permeability  and  losses,  and  a 
measurement  of  the  incremental  permeability  at  various  polarizing  inductions. 

Of  the  several  methods  that  can  be  used  for  measuring  magnetic  properties,  the  ballistic 
ring  test,  due  to  Rowland,  is  perhaps  the  most  reliable.  In  this  test,  a  ring  sample  is  wound 
with  two  uniformly  distributed  windings,  consisting  of  a  primary  connected  to  a  source  of 
current,  and  a  secondary  connected  to  a  ballistic  galvanometer  or  fluxmeter.  The  induction 
produced  by  current  in  the  primary  winding  is  observed  in  terms  of  the  fluxmeter  deflec- 
tion as  the  primary  current  is  changed  suddenly  or  reversed.  The  use  of  a  ring  sample 
eliminates  the  possibility  of  errors  due  to  air  gaps.  To  obtain  uniformity  of  magnetizing 
force  throughout  the  sample,  the  ratio  of  the  outside  diameter  to  the  inside  diameter  of 
the  ring  should  be  not  greater  than  1.2.  Figure  18  shows  a  typical  electrical  circuit  for 


FIG.  18.    Simplified  Diagram  of  a  "Ballistic  Test"  Circuit 


fl  n  ^  including  *  m^tual  Muctance  for  calibrating  the  galvanometer  or 

flurmeter,  <?,  and  resistances  R  and  switches  S  for  regulating  and  changing  the  current  in 
the  primary  winding,  P.    The  field  is  calculated  using  the  relation- 


MEASUREMENT  OF  MAGNETIC  CHARACTERISTICS      2-73 


and  the  induction  using  the  expression : 

3  -  Kd 


ICM  X  108S 
8CN3A, 


in  these  equations  Nf  =  number  of  turns  in  primary  winding. 

/  =  primary  current  in  amperes. 

d  —  mean  diameter  of  ring  in  centimeters. 
Ic  —  calibrating  current  in  amperes. 
M  =  calibrating  mutual  inductance  in  henrys. 

dc  =  calibrating  deflection  resulting  from  a  reversal  of  I0. 
Ns  =  number  of  turns  in  secondary  winding. 
A8  —  sectional  area  of  sample  in  square  centimeters. 

5  —  fiuxmeter  deflection  resulting  from  reversal  of  current  in  primary 
winding. 

Straight  bar  or  rod  samples  are  sometimes  tested  with  this  circuit.  A  long  solenoid  is 
used  for  producing  the  magnetizing  force,  and  the  secondary  winding  or  search  coil  is 
placed  around  a  short  central  portion  of  the  sample.  However,  under  these  circumstances 
the  true  magnetizing  force  is  difficult  to  determine  because  the  field  from  the  magnetic 
poles  produced  at  the  ends  of  the  sample  reacts  with  the  field  of  the  solenoid.  The  field 
created  by  the  sample  itself  is  sometimes  called  the  end  effect  or  demagnetizing  field.  Its 
value  is  usually  specified  by  the  demagnetising  factor,  JV,  which  depends  on  the  ratio 
length/diameter  of  the  rod.  The  field,  H,  acting  at  the  center  of  the  rod  is  the  resultant 
of  the  field  in  the  solenoid,  J?o,  and  the  demagnetizing  field: 

H  =  Ho  -  jj-  (B  -  H) 

The  apparent  permeability,  //,  is  given  by  B/HQ,  and  its  relation  to  the  true  permeabil- 
ity, ju  is  given  by: 

I  =  I  _  ~ 
M  ~~  /*'        4?r 

The  relation  between  p  and  JJL'  for  cylindrical  rods  is  shown  graphically  in  Fig.  19. 

LENGTH 
DIAMETER 
_10    20    40  60  IO2 


IO     20    40  60  To2 


105 


JO3  IO4 

TRUE   PERMEABILITY,  /JL 

FIG.  19.     Relation  of  the  Apparent  Permeability  to  the  True  Permeability  for  Cylindrical  Rods  of 
Various  Ratios,  m,  of  Length  to  Diameter.    Also,  Demagnetizing  Factors,  N/4ir,  as  Dependent  on  m. 


2-74 


PROPERTIES  OF  MATERIALS 


Various  types  of  permeameters  are  also  used  with  the  circuit  shown  in  Fig.  18.  They 
are  especially  useful  for  measuring  permanent-magnet  materials.  Permeameters  usually 
are  designed  to  test  straight  bar  samples  clamped  against  a  yoke  of  very  low  reluctance. 
Measurement  of  samples  of  high  permeability,  so  tested,  are  subject  to  error  due  princi- 
pally to  the  effects  of  the  air  gaps  in  the  circuit.  Only  a  few  of  the  many  types  will  be 
described. 

The  Fahy  permeameter  is  commonly  used  for  testing  materials  like  iron  and  silicon-iron 
as  well  as  some  of  the  magnet  steels  of  relatively  low  coercive  force.  It  is  suitable  for  tests 
at  magnetizing  forces  up  to  300  oersteds.  This  instrument,  shown  in  Fig.  20,  has  one 
large  magnetizing  winding  on  a  yoke  of  silicon-iron.  Pole  pieces  extending  from  either 
end  of  the  yoke  are  arranged  so  that  bar  samples  can  be  clamped  to  them.  The  magnetiz- 
ing force  is  measured  by  an  air-core  solenoid  (H  coil)  mounted  across  the  ends  of  the  pole 
pieces  and  above  the  sample.  A  winding  enclosing  the  sample  acts  as  a  secondary  (B  coil) 
and  measures  the  induction  with  the  aid  of  a  galvanometer  as  in  Fig.  18. 


MAGNETIZING 
•  COIL 


FIG.  20.    Descriptive  Drawing  of  the  Fahy  Perme- 
ameter 


SPECIMEN 


FIG.  21.    Descriptive  Drawing  of  the  Babbitt 
Permeameter 


The  Babbitt  permeameter  (see  Fig.  21)  can  be  used  for  testing  at  magnetizing  forces  as 
high  as  1000  oersteds.  Several  different  high-permeability  materials  are  used  in  the  yoke 
to  give  low  reluctance  over  a  wide  range  of  magnetizing  force,  and  the  magnetizing  coil 
encloses  the  sample  instead  of  being  wound  on  the  yoke  alone  as  in  the  Fahy.  Smaller 
windings  are  placed  on  the  yoke  to  compensate  for  the  reluctance  of  the  air  gaps.  Mag- 
netizing force  and  induction  are  measured  with  H  and  B  coils  placed  near  and  around  the 
middle  of  the  sample. 

More  accurate  tests  of  high-permeability  materials  can  be  made  by  means  of  the  Bur- 
rows permeameter.  This  type  requires  two  samples  clamped  between  two  connecting  yokes 
of  high-permeability  material  completing  the  magnetic  circuit.  In  addition  to  the  mag- 
netizing coils  around  the  samples,  there  are  compensating  coils  around  each  end  of  the 
samples  to  give  more  adequate  corrections  for  the  effect  of  the  air  gaps  at  the  joints.  B 
coils  are  wound  around  the  middle  of  each  sample,  and  two  search  coils  are  placed  on 
either  side  of  each  B  coil.  With  proper  adjustment  the  conditions  of  test  more  nearly 
approach  those  in  the  ring  test,  but  the  test  is  time-consuming  because  of  the  number  of 
adjustments  required. 

Small  air  gaps  are  not  very  important  when  testing  permanent-magnet  alloys,  but 
modern  magnet  materials  require  permeameters  that  can  produce  very  high  magnetizing 
fields.  The  saturation  permeameter  and  the  high-H  permeameter  are  frequently  used  for 
this  purpose. 

The  saturation  permeameter  is  very  similar  to  the  Babbitt  permeameter  except  that  the 
magnetizing  coil  is  larger  and  artificially  cooled,  and  no  compensating  coils  are  used  on 
the  yoke.  Magnetizing  forces  as  high  as  2500  oersteds  are  readily  obtained  with  this 
instrument. 

The  higbrH-permeameter  developed  by  the  Bureau  of  Standards  can  produce  even  more 
powerful  fields  and  can  be  used  to  test  any  of  the  modern  permanent-magnet  alloys.  In 
this  permeameter  four  large  coils  are  used.  Two  are  wound  on  the  yokes  and  two  on  the 
pole  pieces  clamped  to  the  specimen  (see  Fig.  22).  The  induction  is  measured  in  the  usual 
way  with  a  coil  wound  around  the  center  of  the  sample.  A  small  rotatable  H  coil  is  ar- 
ranged so  that  readings  can  be  taken  at  different  distances  from  the  surface  of  the  sample* 
the  data  so  obtained  give  a  curve  of  the  variation  in  H  and  indicate  by  extrapolation  the 
true  value  of  field  at  the  center  of  the  sample. 

In  practice,  magnetic  materials  are  often  subjected  to  alternating  fields,  and  it  is,  there- 
fore, important  to  measure  magnetic  permeability  and  energy  losses  by  a-c  methods 
These  Delude  the  use  of  a-c  bridges,  wattmeters,  cathode-ray  oscilloscopes,  and  various 
other  instruments,  a  few  of  which  will  be  described. 


MEASUREMENT  OF  MAGNETIC   CHARACTERISTICS      2-75 

One  of  the  common  bridge  circuits  is  the  Maxwell  bridge.  The  simplest  form  of  this 
bridge  consists  of  a  pair  of  resistances  for  ratio  arms,  a  variable  resistor  and  variable  in- 
ductance to  balance  the  impedance  of  the  sample,  and  a  detector  which  may  be  a  gal- 
vanometer, a  sensitive  voltmeter,  or  a  telephone  receiver  (see  Fig.  23).  The  bridge  is 
useful  for  measuring  apparent  permeabilities  and  losses  for  low  inductions  at  frequencies 
in  the  audio  range.  It  is  not  suitable  for  testing  at  high  inductions  because  of  errors 
introduced  by  wave-form  distortion. 


FIG.  22.    Schematic  Diagram  of  the  High-H 
Permeameter 


FIG.  23.    Simplified  Maxwell  Bridge  Circuit  for  Deter- 
mining Equivalent  Series  Resistance,  R,  and  Induct- 
ance, L,  of  a  Coil  Containing  a  Specimen  of  Magnetic 
Material 


Incremental  or  superposed  permeability  measurements  can  also  be  made  with  the  Max- 
well bridge,  by  using  an  additional  winding  on  the  sample  connected  in  series  with  a  large 
inductance  and  a  source  of  direct  current.  The  inductance  in  the  d-c  circuit  must  be 
large  enough  to  keep  the  alternating  current  in  this  circuit  at  a  negligible  value.  Incre- 
mental permeability  is  important  in  the  transformers  of  vacuum-tube  amplifiers  and  in 
polarized  apparatus  such  as  telephone  receivers  where  the  material  is  subjected  to  both 
d-c  and  a-c  fields  (see  Fig.  24) . 


REVERSIBLE  PERMEABILITY  ,  fJlr 
p  p  —  .  —  w  |v 
o  £•  o>  to  b>  b  ji 

-^ 

\ 

\  45  PERMALLOY 

\ 

VANA 
PERME 

DIUM 
NDUR 

A 

•\ 

\ 

ARMCO   IRON 

[  

— 

^,\s^ 

=JB«  

•sx 

"0  4  8  13  16  20  24 

POLARIZING  INDUCTION,  B0 

FIG.  24.     Change  in  Reversible  Permeability  with  Polarizing  Induction  for  Several  Materials 


Properties  are  often  determined  at  frequencies  above  the  audio  range  with  a-c  bridges 
of  the  resistor-capacitor  type,  and  Q-meters.  For  further  information  on  a-c  bridge 
methods,  see  Measurement  of  Inductance  and  Effective  Resistance,  Section  11. 

For  magnetic  materials  in  sheet  form,  it  is  convenient  to  test  samples  made  of  sheared 
strips.  The  Epstein  test,  in  common  use  for  testing  the  core  loss  of  materials  such  as  silicon- 
iron  sheet,  uses  samples  of  this  form.  Primary  and  secondary  windings  are  placed  on  four 
hollow  square  forms  mounted  in  the  form  of  a  square.  The  primary  exciting  current  is 
measured  with  an  a-c  ammeter,  A,  and  the  induction  is  indicated  by  an  "average"  volt- 


2-76 


PROPERTIES  OF  MATERIALS 


meter  sometimes  called  a  flux  voltmeter,  connected  across  the  secondary  as  indicated  in 
Pig.  25.  The  core  loss  is  determined  from  the  reading  of  a  wattmeter,  W.  The  test  strips 
used  in"  the  Epstein  are  usually  stacked  with  overlapping  joints  for  permeability  tests  but 
may  be  stacked  with  butt  joints  for  core-loss  tests.  This  method  gives  reliable  data  up  to 


(a) 


v OUTER        INNER,     FORM   STRIPS 
WINDINGS 


FIG.  25.     Descriptive  Drawing  and  Simplified  Circuit  Diagram  for  the  Epstein  Test  for  Determining 

Watt  Loss 

high  inductions,  and  it  is  particularly  useful  for  studying  grain-oriented  materials.  For 
details  of  this  test  and  other  test  methods,  the  latest  issue  of  ASTM  specifications  should 
be  consulted. 

Cathode-ray  oscilloscopes  are  sometimes  used  to  give  rapid  indications  of  the  a-c  prop- 
erties of  materials.  By  means  of  a  simple  integrating  circuit,  hysteresis  loops  can  be 
produced  on  the  screen.  A  simplified  circuit  of  this  type  is  shown  in  Fig.  26.  This  test  is 


SPECIMEN 

FIG.  26.    Simplified  Circuit  for  Producing  Hysteresis  Loops  on  Cathode-ray  Screen 

not^  as  precise  as  those  described  above,  but  because  of  its  rapidity  it  finds  frequent  appli- 
cation in  certain  types  of  production  testing. 


BIBLIOGRAPHY 


Frequency  Cores  of  Hieh  Permeability,  Electronic  Industries, 


ASTM  Spec.  A-34. 

L,  G.  0.,  and  H.  : 

>1.  4,  86  (1945). 

TH,    h.R.M.,  Present  Status  of  Ferromagnetic  Theory,  Elec.  Eng.,  Vol.  54,  1251  (1935). 
The  Physical  Basis  of  Ferromagnetism,  Bell  Sys.  Tech.  /.,  Vol.  19,  1  (1940) 

t&rtL-Qj-       J^         ariA    TUT       Vlrt-MTi.*       J?«~~^™__ J-- /?_     /~\  .  \  '«          .  '      —  V      ..','.    - 


g^ker,  R.,  and  W.  Doring,  Ferromagnetism  (in  German).     Springer,  Berlin  (1939). 
Cioffi,  P.  P.,  Eydrogemzed  Iron,  Phys.  Rev.,  Vol.  39,  363  (1932)  \*-™*)- 

DiUi^rL.i  ?r*'ia?do?n  A&oS°aorth'  Heat  Tr*^nt  of  Magnetic  Materials  in  j 


..       Physics,  Vol.  6,  279  (1935). 


a  Magnetic  Field, 


BIBLIOGRAPHY  2-77 

Electrical  Engineering  Staff  of  M.  I.  T.,  Magnetic  Circuits  and  Transformers.    John  Wiley,  New  York 

(1944). 
Ellis,  W.  C.,  and  E.  E.  Schumacher,  A  Survey  of  Magnetic  Materials  in  Relation  to  Structure,  Bell 

Sys.  Tech.  J.,  Vol.  14,  8  (1935). 

Elmen,  G.  W.,  Magnetic  Alloys  of  Iron,  Nickel  and  Cobalt,  Elec.  Eng.,  Vol.  24,  1292  (1935). 
Hornfeck,  A.  J.,  and  R.  F.  Edgar,  The  Output  and  Optimum  Design  of  Permanent  Magnets  Subjected 

to  Demagnetizing  Forces,  A.I.E.E.  Trans.,  Vol.  50,  1017  (1940). 
Legg,  V.  E.,  Survey  of  Magnetic  Materials  and  Applications  in  Telephone  System,  Bell  Sys.  Tech.  J.t 

Vol.  18,  438  (1939). 
Legg,  V.  E.,  and  F.  J.  Given,  Compressed  Powder  Molybdenum  Permalloy  for  High  Quality  Inductance 

Coils,  Bell  Sys.  Tech.  J.,  Vol.  19,  385  (1940). 

Messkin,  W.  S.,  and  A.  Kussmann,  Ferromagnetische  Legierungen  (in  German).     Springer,  Berlin  (1932). 
Rotors,  H.  C.,  Electromagnetic  Devices.     John  Wiley,  New  York  (1941). 
Ruder,  W.  E.,  Permanent  Magnet  Steels,  Iron  Age,  Vol.  157,  65  (1946). 
Sanford,  R.  L.,  An  Apparatus  for  Magnetic  Testing  at  Magnetizing  Forces  up  to  5000  Oersteds,  Bur. 

Standards  J.  Research,  Vol.  23  (1939). 
Permanent  Magnets,  Bur.  Standards  Circ.  C448  (1944). 

Spooner,  T.,  Properties  and  Testing  of  Magnetic  Materials.    McGraw-Hill,  New  York  (1927). 
Woldman,  N.  E.,  and  R.  J.  Metzler,  Engineering  Alloys.     Am.  Soc.  Metals  (1945). 
Yensen,  T.  D.,  Magnetically  Soft  Materials,  Trans.  Am.  Soc.  Metals,  Vol.  27,  797  (1937). 


SECTION  3 
RESISTORS,  INDUCTORS,  CAPACITORS 


RESISTORS  AND  RHEOSTATS 

BY  PAUL  S.  DARNELL  AND 
ART.  ARTHUR  H.  SCHAPER  PAQE 

1.  General 02 

2.  Wire- wound  Resistors 05 

3.  Composition  Carbon  Resistors 11 

4.  Deposited-carbon  Resistors 15 

5.  Metal  Film  Resistors 17 

6.  Potentiometers  and  Rheostats 17 

7.  Special-purpose  Resistors 21 

VARISTORS  AND  THERMISTORS 

BY  N.  Y.  PRIESSMAN 

8.  Copper-cuprous  Oxide  Varistor 23 

9.  Silicon  Carbide  Varistors 26 

10.  Thermistors 28 

INDUCTORS  WITH  AIR  CORES 

BY  L.  M.  HERSHEY 

11.  Properties  of  Air-core  Inductors 31 

12.  Electrical  Design  Considerations 32 


ART.  PAGE 

13.  Mechanical  Design  Considerations 36 

14.  Inductor  Design  Formulas 38 

FERROUS-CORED  INDUCTORS 
BY  A.  J.  ROHNER 

15.  Low-frequency,  Sheet-core  Inductors ...  42 

16.  High-frequency,    Powdered-core    Induc- 

tors    50 

CAPACITORS 

BY  JAMES  I.  CORNELL 

17.  Classification  of  Capacitors 53 

18.  Variable  and  Adjustable  Capacitors ....  55 

19.  Impregnated-paper  Capacitors 60 

20.  Mica  Capacitors 64 

21.  Ceramic  Dielectric  Capacitors 67 

22.  Electrolytic  Capacitors 68 


3-01 


RESISTORS,  INDUCTORS,  CAPACITORS 
RESISTORS  AND  RHEOSTATS 

By  Paul  S.  Darnell  and  Arthur  H.  Schaf  er 

1.  GENERAL 

DEFINITIONS.  A  resistor  is  a  circuit  element  whose  primary  function  is  to  introduce 
electric  resistance  into  an  electric  circuit. 

A  rheostat  is  an  adjustable  resistor  which  is  provided  with  mechanical  means  for  changing 
its  resistance  value  without  opening  the  circuit  in  which  it  may  be  connected.  Its  primary 
function  is  to  adjust  the  current  in  a  circuit  or  portion  of  a  circuit  in  which  it  is  connected. 
It  may  be  in  the  form  of  a  three-terminal  potentiometer  as  defined  below,  or  it  may  have 
only  two  terminals. 

A  potentiometer  is  defined  by  American  Standards  Association  as  a  measuring  instrument 
by  means  of  which  an  electromotive  force  in  one  of  the  arms  of  the  circuit  may  be  measured 
in  terms  of  one  or  more  other  electromotive  forces  and  the  constants  of  the  potentiometer 
circuit.  However,  the  term  potentiometer  is  most  commonly  used  to  refer  to  any  adjust- 
able resistor  having  three  terminals,  two  of  which  are  connected  to  the  ends  of  the  resist- 
ance element  and  the  third  to  a  contact  which  traverses  the  resistance  element  without 
discontinuity.  Its  primary  function  is  to  convert  an  impressed  voltage  into  a  source  of 
voltage  which  can  be  adjusted  from  a  small  percentage  of  the  impressed  voltage  to  approx- 
imately the  magnitude  of  the  impressed  voltage. 

PHYSICAL  AND  ELECTRICAL  CONSIDERATIONS.  Numerous  factors,  which  will 
determine  the  type  as  well  as  the  physical  size  and  shape,  must  be  considered  in  the  design 
or  selection  of  the  proper  resistor  or  rheostat  for  a  specific  application.  The  most  important 
of  these  are:  (1)  resistance  value  and  tolerance;  (2)  power  dissipation  under  normal  and 
trouble  operating  conditions;  (3)  frequency  characteristic,  phase  angle,  and  change  in 
resistance  over  a  frequency  range;  (4)  stability  with  age  and  changing  conditions  of  tem- 
perature, humidity,  and  voltage;  (5)  mounting  arrangements — space  and  shape  require- 
ments; and  (6)  relative  cost  of  available  resistor  structures  which  will  fulfill  the  circuit 
requirements  in  whole  or  in  part. 

Resistance  Value.  In  general  two  classes  of  material  are  used  for  resistors:  (1)  pure 
metals  and  metal  alloys  in  which  resistance  value  for  any  given  metal  is  determined  largely 
by  its  physical  dimensions,  and  (2)  a  composition  or  mixture  of  a  carbon  or  metallic  con- 
ductor with  an  insulating  material  in  which  the  resistance  value  is  determined  by  the 
relative  proportion  of  conductor  and  insulator. 

In  the  first  group  the  element  is  usually  a  wire  or  strip  having  relatively  low  resistivity, 
which  places  a  definite  limitation  on  the  resistance  value  that  can  be  achieved  for  a  given 
volume.  The  advantages  are  high  stability  with  age,  low  temperature  coefficient,  low 
voltage  coefficient,  and  low  microphonic  noise  level.  Its  disadvantages  are  relatively  high 
cost  for  higher  resistance  values,  corrosion  hazard  under  adverse  conditions  of  voltage  and 
humidity,  limitation  on  high  resistance  values,  and  usually  poor  frequency  characteris- 
tics at  higher  frequencies.  Resistance  range  is  normally  from  about  0.1  ohm  to  1  megohm. 
In  addition  to  the  wire-wound  resistors  in  this  class  are  the  metallic  or  carbon  precision 
film-type  resistors  (to  be  discussed  later)  which  circumvent  some  of  the  disadvantages 
enumerated  here,  since  they  permit  attainment  of  high  resistance  in  small  space  and  have 
greatly  improved  frequency  characteristics. 

In  the  second  group  of  resistors,  since  resistance  value  becomes  a  matter  of  composition 
of  a  mixture,  the  entire  range  of  resistors  in  general  use  (10  ohms  to  22  megohms)  can  be 
made  in  the  same  physical  form  and  volume,  so  that  size  is  determined  largely  by  power 
dissipation  desired.  Other  advantages  are  low  cost,  improved  performance  at  high  fre- 
quency, and,  when  wattage  is  not  a  factor,  small  size,  light  weight,  and  ease  of  mounting. 
Disadvantages  are  lesser  stability  with  time,  temperature,  and  humidity  compared  with 
wire-wound  resistors,  broader  manufacturing  tolerances,  and  higher  noise  level. 

A  third  class  of  resistors,  namely  varistors  and  thermistors  (see  articles  8  to  10),  is 
currently  finding  wide  application  as  special-purpose  circuit  elements  that  are  character- 
ized chiefly  by  their  high  sensitivity  to  temperature  and  voltage  and  their  relatively  high 

3-02 


GENERAL  3-03 

resistivity.    In  this  group  the  elements  consist  of  semiconductor  materials  in  the  range 
between  metallic  conductors  and  insulating  materials. 

Resistance  Tolerance.  The  matter  of  manufacturing  tolerance  with  respect  to  the 
limits  of  resistance  value  within  which  a  resistor  shall  initially  be  held  is  important  from 
the  standpoint  both  of  the  proper  functioning  of  the  circuit  involved  and  of  the  design  of 
the  resistor  itself.  Practical  tolerances  of  low-cost  resistors  commercially  available  are 
of  the  order  of  ±5  per  cent  to  ±20  per  cent  for  composition  type,  and  ±1  per  cent  to 
±10  per  cent  for  wire-wound  and  precision  film  types. 

Of  equal  importance  to  the  initial  manufacturing  tolerances  is  the  desired  stability  of 
the  resistor  during  its  life.  To  obtain  stability  of  a  lesser  order  of  magnitude  than  the 
original  manufacturing  limits  can  be  fairly  simple  at  even  the  closest  limits  of  the  ranges 
indicated  above  but  will  probably  become  the  major  problem  or  even  an  impossibility  for 
units  adjusted  to  initial  tolerances  of  one-quarter  to  one-tenth  of  those  figures,  for  example. 

ENERGY  DISSIPATION—TEMPERATURE  RISE— POWER  RATINGS.  Since  nor- 
mally the  total  electrical  energy  supplied  to  a  resistor  is  dissipated  in  the  form  of  heat, 
the  resultant  temperature  rise  will,  under  adverse  conditions,  constitute  a  potential 
hazard  both  to  the  resistor  itself  and  to  the  materials  of  surrounding  objects.  Industrial 
Control  Standards  (NEMA,  July  1,  1946,  IC4-22)  state  that  the  temperature  rise  for  bare 
resistive  conductors  shall  not  exceed  375  deg  cent  as  measured  by  a  thermocouple  in  con- 
tact with  the  hottest  spot  on  the  bare  conductor,  and  for  resistive  conductors  imbedded 
shall  not  exceed  300  deg  cent  as  measured  by  a  thermocouple  in  contact  with  the  hottest 
spot  on  the  surface  of  the  imbedding  material.  The  establishment  of  such  maximum  power 
ratings  is  possible  only  for  power-type  resistors  constructed  entirely  of  inorganic  materials 
not  adversely  affected  either  physically  or  chemically  by  the  heat  generated.  Ratings 
for  other  resistors  are  normally  established  on  the  basis  of  the  maximum  temperature  at 
which  they  can  operate  continuously  without  deterioration  of  their  performance  or  their 
component  parts.  Since  power  ratings  under  the  various  standards  are  predicated  on 
operation  in  still  air  and  free  space  at  an  ambient  temperature  of  the  order  of  25  deg  cent,  a 
status  seldom  realized  under  actual  operating  conditions,  good  engineering  practice  dictates 
derating  the  resistor  to  reflect  the  specific  conditions  of  use.  In  the  absence  of  exact  data 
to  indicate  the  amount  of  derating  necessary,  a  figure  of  50  per  cent  is  commonly  applied. 
In  addition  to  consideration  of  the  normal  wattage  at  which  the  resistor  will  be  required 
to  operate,  it  is  frequently  advisable  to  determine  the  power  the  resistor  must  dissipate 
under  predictable  abnormal  or  trouble  conditions,  and  when  possible,  to  select  a  resistor 
that  will  operate  safely  under  such  conditions  as  well. 

FREQUENCY  CHARACTERISTIC.    In  considering  the  behavior  of  a  resistor  over  a 
frequency  range,  it  must  be  remembered  that  in  any  practical  design  in  which  the  element 
has  finite  dimensions  it  will  necessarily  contain  components 
of  all  three  parameters  (Fig.  1 — R,  L,  and  C)  and  will 


approach  the  characteristics  of  a  pure  resistor  over  a  limited  — ?— vWW — ^^fi^ — j — • 
frequency  range.     The  problem  of  design  is  to  approach  a 

pure  resistance  with  the  inductance  and  capacitance  re-         '  |[ I 

garded  as  parasitic  values,  useful  only  to  the  extent  to  Q 

which  they  can  be  made  to  nullify  each  other  over  the    ~      ,      ,.,     .    ,     .  ,T  .       ,      , 
,  .         i.  r    jr  -J.-J.T-  j       A     FIG.  1.     Equivalent  Network  of 

frequency  range  in  which  the  resistor  is  to  be  used.     A    a   Resistor   Having    Inductance 
typical  resistor  may  be  represented  as  a  two-terminal  net-       and  Distributed  Capacitance 
work  comprising  three  elements  as  shown  in  Fig.  1. 

The  impedance  Z  of  this  circuit  may  be  expressed  in  terms  of  effective  inductance  L' 
and  effective  resistance  R'  of  the  system  as  follows:  Z  =  R'  +  juL'.  It  can  readily  be 
shown  that  in  terms  R,  L,  and  C  the  following  relations  hold : 

7?/  .  ?L m 

1  -  co2C(2L  -  CR2)  + 

L-CR2- 


1  -  a?C"(ZL  -  CR2)  +  oj^L2  (2) 

Also  the  phase  angle  &  =  tan^oxLY-SIOr  from  which 

Tan*=a(L-C^-'°2CL2) 

Since  in  a  resistor  both  L  and  C  are  small,  eqs.  (1),  (2),  and  (3)  may  be  rewritten: 

Rf  =  B[l  +  co2C(2L  -  CR2)]  (4) 

L'  -  L  -  CR2  (5) 

Tan  $  = 


3-04  RESISTORS,   INDUCTORS,   CAPACITORS 

It  will  be  observed  that  the  phase  angle  will  be  zero  if  L  —  CW-  —  0,  but  that  the  change 
in  resistance  will  be  zero  only  if  C  =  0  or  if  2L  -  CR2  =  0.  It  follows  then  that  both  the 
phase  angle  and  the  change  in  resistance  with  frequency  can  be  zero  simultaneously  only 
for  the  conditions  C  —  0,  L  -  0.  These  are  important  relations,  for  it  means  that  a 
resistor  may  be  designed  with  a  small  or  zero  phase  angle  over  a  considerable  frequency 
range  merely  by  keeping  to  a  minimum  the  quantities  L/R  and  CR  and  making  them  as 
nearly  equal  as  is  warranted  for  the  particular  design  under  consideration. 

A  second  factor  causing  change  of  resistance  with  frequency  is  the  so-called  skin  effect, 
which  arises  from  the  fact  that  elements  or  filaments  of  current  at  different  points  in  the 
cross-section  of  the  conductor  encounter  different  components  of  inductance.  This  is 
because  of  the  greater  mutual  inductance  between  elements  at  the  center  of  the  conductor. 
These  unequal  inductances  over  the  cross-section  tend  to  produce  unequal  current  densities, 
with  the  minimum  at  the  axis  and  the  maximum  at  the  periphery.  Such  unequal  current 
densities  reduce  the  effective  cross-section  of  the  conductor  and  tend  to  increase  its  effective 
resistance.  Since  the  determination  of  skin  effect  by  computation  is  involved,  it  is  suffi- 
cient to  observe  that  this  effect  may  be  kept  to  the  minimum  by  (1)  using  a  conductor  of 
small  cross-section,  (2)  using  a  conductor  of  high  specific  resistance,  and  (3)  for  a  wire- 
Tvound  resistor,  winding  the  wire  to  have  the  nainimum  effective  inductance.  It  will  be 
seen  that  these  are  also  conditions  for  other  good  characteristics  in  an  alternating-current 
resistor.  (For  a  more  detailed  description  of  skin  effect  and  methods  of  computation,  see 
National  Bureau  of  Standards  Circular  74.) 

Small  Phase  Angle  over  a  Frequency  Range.  As  stated  previously,  to  insure  small  phase 
angle  over  a  given  frequency  range  requires  that  residual  inductance  and  capacitance  be 
held  to  the  minimum,  zero  phase  angle  being  realized  when  one  is  made  to  compensate 
the  other,  Such  compensation  is  achieved  by  satisfying  the  equation  L  —  CR*.  The 
major  problem  for  low  resistance  is  the  reduction  of  inductance;  for  high  resistance  it  is 
the  reduction  of  capacitance.  In  actual  practice  it  is  found  that  for  values  up  to  about 
100  ohms  the  problem  is  one  of  inductance;  for  values  over  several  thousand  ohms  the 
problem  is  mostly  capacitance;  for  values  in  between,  both  capacitance  and  inductance 
must  be  considered.  Of  course  these  figures  show  merely  the  broad  boundaries,  and  con- 
siderable variation  will  be  encountered  due  to  differences  in  structure,  type  of  winding, 
and  capacitance  to  ground. 

STABILITY  "WITH  AGE,  TEMPERATURE,  AND  HUMIDITY.  Aging  may  be  defined 
as  any  permanent  change  in  resistance  value  with  time,  measured  under  the  same  condi- 
tions of  temperature  and  humidity.  It  is  generally  caused  by  strains  set  up  in  the  resistor 
element,  either  in  the  original  manufacturing  process  as  when  wire  is  wound  tightly  on  a 
core  or  when  the  element  is  pressed  or  molded  into  shape,  or  during  the  life  of  the  unit 
when  the  supporting  structure  is  distorted  as  the  result  of  aging  of  the  structure  itself. 
In  a  composition  resistor  it  is  also  brought  about  by  chemical  and  physical  changes  in  the 
materials  of  composition,  particularly  in  the  deterioration  of  the  insulating  materials  used 
in  the  binder  and  filler.  Strains  in  the  winding  may  often  be  relieved  by  a  preaging  treat- 
ment (generally  baking  from  1  to  24  hours).  To  prevent  warpage  of  the  structure,  and 
to  guard  against  other  effects  discussed  later,  resistors  are  protected  against  moisture  in 
various  ways.  Impregnation  with  wax,  varnish,  or  asphalt  compounds  is  common  practice, 
and  frequently  an  exterior  covering  of  moisture-resistant  material  is  used.  Proper  selec- 
tion of  structural  materials  is  probably  the  most  important  single  factor  in  aging  protection. 
Metal,  which  appears  to  be  the  ideal  material  for  this  purpose,  usually  cannot  be  used, 
partly  because  of  its  effect  on  the  residuals,  and  partly  because  proximity  to  the  winding 
would  be  an  added  hazard  in  promoting  electrolytic  corrosion. 

Temperature  Variations.  Stability  with  change  in  temperature  is  obtained  by  a  proper 
selection  of  the  resistive  material.  For  composition-type  resistors  this  material  is  at  present 
limited  almost  entirely  to  carbon  or  graphite  for  the  conducting  portion  of  the  mix;  a 
wide  variety  of  insulating  materials  is  used  for  binder  and  filler,  and  frequently  the  choice 
is  controlled  by  considerations  other  than  temperature  coefficient.  For  wire-wound  resis- 
tors the  desired  temperature  characteristics  are  determined  largely  by  the  wire  selected. 

Humidity  Conditions.  In  addition  to  warping  of  structural  materials  two  other  humidity 
effects  must  be  considered  in  resistance  design:  the  effect  on  residual  inductance  and 
capacitance,  and  the  effect  on  corrosion.  When  moisture  gets  into  a  resistance  structure 
it  causes  a  change  in  the  dielectric  properties  of  the  materials  used  therein,  and  a  conse- 
quent change  in  the  residual  capacitance.  This  will  often  change  the  phase  angle  by  an 
appreciable  amount.  The  danger  of  corrosion  which  takes  place  when  impurities  and 
moisture  are  present  is  generally  the  more  serious  consideration  in  protection  against 
humidity,  especially  where  fine  wires  are  used. 

CLASSIFICATIONS.  Resistors  have  been  variously  classified  in  terms  of  performance, 
power  dissipation,  structure,  usage,  and  manufacturing  tolerances.  Broadly,  the  following 


WIRE-WOUND  RESISTORS 


3-05 


classifications  are  in  general  use  today  and  have  been  adopted,  with  proper  subdivisions, 
by  the  various  standards  agencies:  (1)  fixed  wire-wound,  (2)  fixed  composition  type, 
(3)  precision  film  type,  (4)  rheostats  and  potentiometers  (wire-wound  and  composition), 
(5)  special-purpose  types  (chiefly  thermistors  and  varistors).  Each  of  these  categories 
will  be  treated  separately  below. 

RESISTOR  TESTS  AND  SPECIFICATIONS.  Aside  from  a  marked  quality  improve- 
ment of  the  resistor  product,  one  of  the  major  benefits  derived  from  the  past  war  was  the 
promulgation  and  adoption  by  the  armed  services,  and  to  a  lesser  extent  by  industry,  of 
resistor  standards  of  performance,  size,  resistance  value,  and  tolerance.  These  can  be 
found  by  reference  to  the  following  specifications: 

JAN-R-11  Resistors,  Fixed  Composition. 

JAN-R-26  Resistors,  Fixed  Wire- Wound,  Power  Type. 

JAN-R-93  Resistors,  Accurate,  Fixed,  Wire-Wound. 

JAN-R-184  Resistors,  Fixed,  Wire- Wound  (Low-power). 

JAN-R-22  Rheostats,  Wire- Wound,  Power  Type. 

JAN-R-19  Resistors,  Variable,  Wire-Wound  (Low  Operating  Temperature). 

Specifications  include  tests  of  both  mechanical  and  electrical  qualities,  including  meas- 
urement of  strength  of  leads,  strength  of  resistor,  resistance  load  life,  load  characteristics, 
voltage  characteristic,  temperature  coefficient,  microphonic  noise,  and  effects  of  humidity, 
overload,  and  aging.  In  addition  the  Radio  Manufacturers  Association  is  in  the  process 
of  preparing  corresponding  specifications  for  commercial  usage,  less  severe  and  exacting. 
(See  also  Section  11.) 

2.  WIRE-WOUND  RESISTORS 

The  many  varieties  of  wire-wound  resistors  in  use  are  classified  in  numerous  ways, 
some  by  the  manufacturer  and  some  by  the  application  to  which  the  resistor  is  put.  Thus 
we  have  power  type,  high-frequency  resistors,  precision  type,  flat  type,  tubular,  plate, 
ballast,  cord  type,  lead  mounting,  ferrule  type,  spool  type,  sectionalized  bobbin  type,  and 
many  others.  As  might  be  expected  there  is  a  great  deal  of  overlapping  in  categories,  and 
the  descriptive  terms  are  of  course  relative  and  at  times  misleading.  Wire-wound  resistors 
will  be  arbitrarily  subdivided  here  into  the  several  main  classifications  currently  accepted 
by  most  manufacturers  and  users. 

POWER-TYPE  RESISTORS.  Power  resistors  consist  of  that  class  of  resistor  whose 
primary  function  is  to  dissipate  relatively  large  amounts  of  power  in  a  comparatively 
small  space.  In  this  group,  materials  of  construction  are  selected  first  because  of  their 
ability  to  withstand  heat.  Normally  temperature  rise  is  limited  to  300  deg  cent  which 
with  25  deg  cent  ambient  means  a  hot-spot  temperature  of  325  deg  cent.  Where  organic 
materials  are  used  (usually  to  obtain  increased  resistance  to  humidity  or  to  facilitate 
construction  of  special  features  requiring  machined  or  molded  parts)  temperature  rise  is 
limited  to  125  deg  cent  and  wattage 
ratings  are  reduced  to  about  one-third 
of  maximum  values  obtainable  with  com- 
plete inorganic  construction.  Typical 
construction  consists  of  a  cylindrical 
ceramic-core  tube  with  encircling  band 
terminals  at  each  end,  with  a  single 
inductively  wound  layer  of  wire  made 
of  resistance  material  selected  for  the 
qualities  desired.  Because  of  corrosion 
hazards  and  considerations  of  mechanical 
strength,  choice  of  wire  size  is  usually 
limited  to  minimum  0.002-in.  diameter 
(0.0025  in.  in  JAN  specifications),  al- 
though most  manufacturers  will  wind 
wire  down  to  0.001  in.  or  less  in  diameter 
if  requested  to  do  so.  The  resistor  wind- 
ing is  protected  against  mechanical  injury 
and  the  effects  of  moisture.  Electrical  insulation  is  obtained  by  applying  a  cover  coat  of 
either  vitreous  enamel  or  cement.  A  typical  assortment  of  sizes  commercially  available 
in  this  type  is  shown  in  Table  1. 

Maximum  resistance  value  is  based  on  use  of  0.002-in  .-diameter  wire.  With  0.001-in. 
wire,  values  up  to  eight  times  these  maximum  figures  may  be  obtained.  Wattage  ratings 
are  based  on  a  permissible  300  deg  cent  temperature  rise.  Maximum  voltage  which  may 


Table  1.    Ratings  and  Dimensions 


Nominal 
Ratings, 

Range  of 
Resistance 

Approximate  Core 

Sizes,  in. 

watts 

Values,  ohms 

Diameter 

Length 

5 

0.5  to      1,600 

5/16 

1 

10 

0.3to      5,000 

5/16 

13/4 

20 

0.3to     10,000 

9/16 

2 

25 

0.3  to    13,000 

5/8 

2 

30 

0.5  to    23,000 

3/4 

3 

40 

0.6  to    28,000 

3/4 

31/2 

50 

0.8  to    40,000 

3/4 

41/2 

75 

1.2  to    50,000 

4  1/4 

115 

1.9to    90,000 

11/8 

61/2 

160 

2.  6  to  120,000 

U/8 

81/2 

200 

3.  6  to  150,000 

H/8 

101/2 

3-06 


RESISTORS,   INDUCTORS,    CAPACITORS 


be  applied  is  properly  held  within  about  500  volts  per  inch  of  length,  subject,  of  course, 
to  permissible  power  dissipation. 

Other  varieties  of  power-type  resistors  include  the  following:  strip  type,  plaque  type, 
disk  type,   plate  type,   supported  ribbon,   ferrule  or  axial  terminal,   adjustable  type, 


0.5        i         2 

Frequency  in  megacycles 

FIG.  2.     Resistance  Frequency  Characteristics  of  Several  Wire-wound  Vitreous  Enamel  Resistors — 
50  Watts,  9/i6  ia.  Diameter  x  4  in.  Core  (Courtesy  Ohmite  Mfg.  Co.) 


0.5          1  2 

Frequency  in  megacycles 

FIG.  3.     Reactance  Frequency  Characteristics  of  Several  Wire-wound  Vitreous  Enamel  Resistors — 
50  Watts,  9/16  in-  Diameter  x  4  in.  Core  (Courtesy  Ohmite  Mfg.  Co.) 

molded  type,  and  those  with  multiple  or  tapped  windings.     The  adjustable  type  has  a 
lengthwise  strip  of  winding  bared  down  one  side  of  the  core  tube,  and  the  resistor  is 

provided  with  an  extra  encircling  band  terminal  for  clamp- 
ing down  on  and  tapping  the  winding  at  any  point  along 
its  length. 

Close-tolerance  Resistors.  Although  usual  manufac- 
turing tolerances  available  are  either  ±5  or  ±10  per 
cent,  closer  tolerances  down  to  about  ±1  per  cent  can  usu- 
ally be  obtained  at  increased  cost.  Tolerances  closer  than 
1  per  cent  are  not  generally  furnished  in  power-type  resis- 
tors since  self-heating  frequently  changes  the  resistance 
value  by  at  least  1  per  cent.  In  the  case  of  vitreous 
enamel  resistors  the  high  firing  temperatures  involved  in 
applying  the  protective  coating  and  the  consequent  resist- 
ance changes  during  manufacture  make  closer  adjustment 
impracticable.  Sometimes  varnishes,  lacquers,  or  asphalt 
coatings  are  substituted  for  the  high-temperature  coverings 
to  facilitate  initial  adjustment,  to  permit  use  of  low-tem- 
per attire-coefficient  wires  without  loss  of  desirable  charac- 


Per  cent  rated  load 

»-»  K>  U>-fkOl<T»-vlOOl0C 
OOO  OOOOOOOc 

^ 

s^ 

\j 

**" 

v^ 

\ 

^- 

) 

\ 

^ 

^^ 

\ 

r-\ 

^ 

^ 

\ 

\ 

\ 

20 


40        60       80       100 
Ambient  temperature  in 
degrees  centigrade 


T?T^     A      T>^-  +~      T*       +•         f 
JiG.    4.     ixesistor    Iterating    for 

High  Ambient  Temperatures  as 

Specified  in  JAN-R-26.    Curve   teristics  brought  about  through  additional  heat  treatment, 

d^Sto£$5££l  Cu24e   and-to  pr°^e  furthfr  P'"°»  ^iaSt  weathering  in 
2 — Nominal  rating  based  on  125    service.     Such  special  coatings  may  necessitate  derating 
deg  cent  temperature  rise.          as  much  as  SO  or  90  per  cent. 


WIRE-WOUND  RESISTORS 


3-07 


Layer  Windings.  Power-type  resistors  are  occasionally  wound  in  layers  to  achieve  a 
greater  range  of  resistance  value.  Resistance  wire  having  insulation  capable  of  with- 
standing high  temperatures  as  well  as 
having  a  high  degree  of  mechanical 
strength  must  be  chosen  for  this  type  of 
resistance  winding. 

Non-inductive  Windings.  Many  of  the 
power-type  resistors  may  be  obtained  with 
windings  of  low  inductance.  Generally 
the  Ayrton-Perry  winding  is  used  (see  de- 
scription under  "Precision-type  Resis- 


1400 
1300 


1200 
1100 


l  000 


2    900 
1    800 

_0> 

»    700 

'w 

•S    600 

|    500 

£    400 

300 

200 

100 


2k 


^ 


Pef  cent  of  nominal  watts  a?  sea  level 

SIO  C 

o  c 

\ 

\. 

\ 

\ 

\ 

\ 

\ 

S 

\ 

X 

^Vl 

•v^ 

3                10,000            20,000            30,( 

Altitude  in  feet  above  sea  level 

FIG.   6.     Resistor  Derating  When  Operated 

Above  Sea  Level  (Courtesy  Ward  Leonard 

Electric  Co.) 


220 


160 


120 
100 


20 


I          |          | 

100°  C  rise  curve; 
(cpi 


0      2      4      6      8    10    12    14    16    18   20    22 
Number  of  resistors  in  group 

FIG.  5.  Resistor  Derating  When  Mounted  in 
Groups.  Spacings  are  centerline  to  centerline  of 
resistors  mounted  horizontally — 11/8  in-  diameter 
core  tubes.  All  resistors  of  equal  length  and  power 
rating.  (Courtesy  Ward  Leonard  Electric  Co.) 

tors"),  which  has  the  added  advantage  of  low  distributed  capacitance  and  greatly  in- 
creases the  frequency  range  over  which  the  resistor  can  be  used  effectively,  particularly 
for  low  resistance  values  (up  to  about  10,000  ohms).  Figures  2  and  3  show  plots  of  the 
normalized  effective  series  resistance  and  normalized  effective  series  reactance  against 
frequency  for  both  standard  and  non-induc- 
tive (Ayrton-Perry)  windings  on  50-watt  core 
sizes  for  resistance  values  of  100,  1000,  5000 
ohms,  and  are  representative  of  the  improved 
performance  that  may  be  expected  from  this 
type  of  winding.  Normalized  resistance  or 
reactance  is  to  be  considered  as  the  ratio  of 
resistance  or  reactance  to  the  d-c  resistance  of 
the  resistor. 

Typical  behavior  and  characteristics  of  some 
power-type  resistors  are  shown  in  Figs.  4  to  10. 
Figures  4  to  7  show  the  required  derating  of 
resistors  operated  under  unfavorable  condi- 
tions; Figs.  8,  9,  and  10  show  behavior  under 
overload  and  intermittent  duty. 

LOW-POWER  AND  PRECISION-TYPE 
RESISTORS.  This  category  is  made  up  of 
those  resistors  in  which  power  dissipation  is  a 
minor  consideration;  it  comprises  a  wide  va- 
riety of  shapes  and  sizes.  Since  power  dissipa- 
tion is  not  an  important  factor,  many  more  ma- 
terials are  available  for  body  structures  and 
protective  coatings,  including  plastics  and 
molding  compounds,  which  are  more  readily 
adaptable  to  the  form  of  unit  required. 

Precision  resistors  fall  within  this  category 
and  are  Usually  considered  to  be  resistors  which 
are  manufactured  to  tolerances  of  ±1  per  cent 
or  less  and  which  are  stable  during  their  normal 
life  and  operating  conditions  to  within  toler- 


/250' 


ie  C  rise 


W0     10   20    30    40    50    60    70    80   90  100 
Watts  in  per  cent  of  nominal  continuous  rating 

FIG.  7.  Temperature  Rise  of  Resistors  in 
Free  Air,  and  Enclosed  in  a  Metal  Box.  Unit 
mounted  in  unventilated  metal  box  reaches 
maximum  permissible  .temperature  when  dis- 
sipating only  about  60  per  cent  of  normal  rat- 
ing. (Courtesy  Ward  Leonard  Electric  Co.) 


3-08 


RESISTORS,  INDUCTORS,   CAPACITORS 


300 


OL     •£  rf  <^coi^<£ifl    ^ 
£    \  I  /  //  /        I 


£100 


FIG.  8.  Heating  Time  Required 
to  Raise"  Hot-spot  Temperature 
300  Deg  Cent  for  50-watt  (9/16 
in.x4:  in.)  Resistor  for  Overloads 
up  to  1000  Per  Cent  of  Rated 
Load.  A  100  per  cent  load  will 
bring  full  300  deg  cent  rise  in 
about  10  minutes.  (Courtesy 
Ohmite  Mfg.  Co.) 


20 


40 


60          80         100 
Time  in  seconds 


120        140 


150 


FIG.  9.  Heating  Time  Required  to  Raise  Hot-spot 
Temperature  180,  240,  and  300  Deg  Cent  for  Over- 
loads up  to  1000  Per  Cent  of  Rated  Load.  Based 
on  rating  of  50  watts  for  9/ig  in.  x.  4  in.  vitreous 
enamel  resistor.  Will  reach  300  deg  cent  with  con- 
tinuous operation  at  rated  load.  (Courtesy  Ohmite 
Mfg.  Co.) 


200       300    400500 
Per  cent  of  rated  load 


1000 


1000 

900 

\ 

N^ 

800 

X  / 

V 

^J 

\. 

"°  600 

\% 

-  500 

> 

& 

s*& 

•a  DUU 
£  400 

^\ 

\ 

^ 

•g  wu 
*S  300 

'tf 

% 

ft 

V 

s 

ft 

«. 

"  200 

t 

\ 

\ 
s, 

o 
a. 

100 

\ 

0.5  1  2 

Duty  cycle  in  percent 

10  20      30    40  50     70    100           200 

Maximum  permissible  heating  time  in  seconds 


3      4    5  6  78  10 


FIG.  10.  Permissible  Duty- 
cycle  for  50-watt  (9/ie  in.  x 
4  in.)  Resistor  Based  on  a 
Maximum  Hot-spot  Tempera- 
ture Rise  of  300  Deg  Cent. 
Curve  A — Per  cent  of  rated 
load  at  which  resistor  may  be 
operated  on  intermittent  duty. 
Curve  B — Maximum  time  re- 
sistor may  operate  during  any 
one  cycle.  (Courtesy  Ohmite 
Mfg.  Co.) 


WIRE-WOUND  RESISTORS 


3-09 


ances  of  something  less  than  the  initial  adjustment,    A  few  of  the  types  in  general  use 
today  are  the  following: 

Spool  Type,  in  which  the  winding  is  applied  to  a  core  body  in  the  form  of  a  spool,  either 
molded  or  fabricated,  to  the  dimensions  necessary  to  accommodate  sufficient  wire  for 
maximum  resistance  value  desired  and  to  dissipate  rated  watts  with  the  minimum  of 
self-heating.  The  spool  may  be  divided  into  sections  to  control  the  voltage  gradient 
between  individual  turns  and  between  sections  of  winding  and  to  permit  some  improve- 
ment in  frequency  characteristic  over  a  winding  applied  in  a  single  section.  Typical  sizes 
available  are  shown  in  the  tabulation. 


Spool  Dimensions, 
in. 

Watts 
Rating 

Maximum  Resistance 
Value,  megohms 
Minimum  0.0015 
in.-diameter  Wire 

Diameter 

Length 

19/32 
19/32 
25/32 

19/32 
H/32 
19/32 

0.5 
0.75 
1.0 

0.15 
0.4 
1.0 

Flat-type  resistors  (card-type)  usually  consist  of  a  single  layer  of  bare  or  insulated  wire, 
wound  on  a  flat  card  of  insulating  material.  The  wound  card  is  either  encased  in  molding 
compound  or  is  given  a  protective  coating  of  lacquer  or  varnish.  Maximum  resistance 
value  is  typically  of  the  order  of  5000  ohms  per  inch  of  length  and  wattage  dissipation 
about  1  watt  per  square 
inch  of  radiating  surface.  Table  2.  Net  Residuals  of  Typical  Resistor  Windings 

Pad-type  resistors  con- 
sist of  multiple  windings 
on  a  spool  or  card  to  form 
resistor  networks  suitable 
for  use  as  attenuators. 

Plug-in  type  resistors 
are  provided  with  termi- 
nations at  one  end  in  the 
form  of  a  vacuum-tube 
base,  providing  means  for 
inserting  attenuator  pads 
in  a  circuit. 

Flexible  resistors  are 
made  in  the  form  of  a  helix 
wound  on  a  flexible  insu- 
lating cord  which  in  turn 
is  insulated  with  a  flexible 
sleeving. 

WINDING  TYPES. 
Numerous  types  of  wind- 
ing are  commonly  used  in 
resistors  in  an  effort  to  at- 
tain low  residuals  of  ca- 
pacitance and  inductance 
or  simply  to  achieve  the 
desired  resistance  value  in 
the  most  economical  way. 
Some  of  these  winding 
types  are  shown  here. 
Typical  residuals  obtained 
are  shown  in  Table  2. 

Continuous  Winding. 
The  simplest  of  all  wind- 
ings is  one  in  which  the 
wire  is  wound  on  a  core 
continuously  in  one  direc- 
tion, known  generally  as  a 
continuous  or  inductive 
winding.  If  the  wire  is  wound  in  layers,  the  residuals  of  both  capacitance  and  inductance 
are  large  and  the  resistor  can  be  used  merely  for  very  low  frequencies  or  direct-current 
applications.  By  winding  in  a  single  layer  on  a  core  of  small  cross-sectional  area  such  as  a 


Winding  Type 

Resistance, 
ohms 

Approximate 
Net  Residual 

Continuous  (flat-card  type) 

0-4.5 
4.5-20 
20-600 
1,000 
10,000 

0-1.  5  ith 
0.5-3.  5  /ih 
3.5-15/xh 
70  Mh 
±3Mjuf 

Continuous  (tubular  power  type) 

0-     200 
200-     800 
800-2,000 
2,000-5,000 

2-1  5  ^h 
15-60  juh 
60-  130  jit 
1  30-500  /*h 

Bifilar 

10 
100 
1,000 
3,500 

0.5  Mb 
1.5Mb 
100  MM£ 
300  wif 

Reverse  layer 

1,000 
3,500 
10,000 

10  wrf 
30/tMf 
lOOwtf 

Reverse  section 

1,000 
3,500 
10,000 

100  A*h 
300  juh 
±5  jtt/xf 

Reverse  half-section 

1,000 
3,500 
10,000 

d=5wrf 

±5  /i/rf 
30  ftfd 

Mandrelated  filament 

'1,000 
3,500 
10,000 
35,000 
100,000 
200,000 

1-2  «rf 
1-2  w£ 
1-3  wd 
2-4  wf 
2-5  «uf 
3-8  w*f 

Parallel  opposing  (Ayrton-Perry) 

100 
1,000 
10,000 

0-1  Mh 
1-2  /*h 
1-2  pid 

3-10 


RESISTORS,   INDUCTORS,    CAPACITORS 


flat  card  or  small-diameter  core,  the  inductance  and  capacitance  are  reduced.  By  spacing 
the  turns  and  keeping  the  wire  size  to  the  minimum  almost  any  desired  residual  may  be 
achieved  with  the  obvious  limitation  that  short  lengths  of  wire  become  increasingly  difficult 
to  adjust. 

Bifilar  Winding.  This  winding  is  used  most  generally  where  low  inductance  is  required. 
In  this  winding  the  wire  is  bent  back  on  itself  at  the  midpoint  so  that  the  two  half-lengths 
are  side  by  side,  separated  by  the  insulation  only.  In  this  straight  form  any  given  wire 
has  the  minimum  possible  inductance.  This  minimum  inductance  L  may  be  computed 
from  the  formula : 

L  =  0.005Z(2.303  logio  ^r  +  0.25)  ph 


Loop  or  splice 
at  midpoint  tied 
down  to  core 


where  I  —  total  length  of  wire  in  inches,  D  —  distance  apart,  between  centers,  d  =  diam- 
eter of  the  bare  wire,  and  juh  =  microhenries.  D  and  d  are  expressed  in  the  same  units. 
In  practice  this  loop  is  generally  wound  on  a  core  (see  Fig.  11), 
often  in  several  layers,  with  the  result  that  L  is  increased  slightly. 
Spool  The  bifilar  winding  has  the  disadvantage  of  high  distributed  capac- 
itance. Next  to  a  straight  inductive  winding,  it  is  the  cheapest 
winding  to  apply. 

A  modification  of  the  straight  bifilar  winding  is  the  series-bifilar 
winding  in  which  the  total  length  of  wire  is  divided  into  sections, 
each  wound  separately  as  a  bifilar  loop  and  the  sections  connected 
in  series.  This  has  the  advantage  of  maintaining  minimum  induct- 
ance and  of  reducing  the  capacitance  by  any  desired  amount, 
inasmuch  as  the  capacitance  is  approximately  inversely  propor- 
tional to  the  square  of  the  number  of  sections.  This  results  in 
a  good  resistor,  but  it  is  expensive  to  apply,  and  since  simpler 
windings  generally  give  practically  the  same  results  it  is  seldom 
used  in  regular  production. 
Reverse  Layer  Winding.  The  reverse  layer  winding  (see  Fig.  12)  is  used  where  the 
frequency  range  is  low  enough  to  permit  appreciable  residuals  and  where  the  spool  is  not 
divided  into  sections  for  a  sectional  winding.  The  direction  of  turns  is  reversed  for  each 
layer.  It  approximates  the  series-bifilar  winding,  but,  because  each  layer  must  be  secured 
in  place  before  reversing  turns,  it  is  expensive  where  a  large  number  of  layers  are  applied. 
To  be  non-inductive  it  must  have  an  even  number  of  full  as  well  as  partial  layers. 

Spool^ 


Two  wires  wound 
together  in  parallel 
for  one  or  more  layers 


FIG.  11. 


Bifilar  Wind- 
ing* 


- 


Section  through  winding  on 

center  line  of  spool 
Notes: 

1.  Section  shows  alternate  fufl 
layers  and  finaJ  partial  layers 
wound  in  opposite  dlrectians- 

O  indicates  layer  wound 
towards  observer.  0  indicates 
layer  wound  away  from  obser* 
ver. 

2.  Winding  always  consists  of 
even  number  of  fuU  and 
partial  layers. 

FIG.  12.    Reverse  Layer  Winding 


m 


~^~-~,  .-^.w*  ..JJ 

B  111  mi 


Section  through  winding  on  center  line  of  spool 
Notes: 

1.  ©  indicates  section  of  winding  wound  toward  the 
observer. 

«  indicates  section  of  winding  wound  away  from 
observer. 

2.  Approximately  the  same  numberof  turns  are  wound 
in  each  of  the  sections. 

3*  Figures  indicate  winding  order. 

FIG.  13.     Reverse  Section  Winding 


Reverse  Section  Winding.  The  reverse  section  winding  (see  Fig.  13)  is  used  largely 
for  high  resistance  values.  This  winding  is  divided  into  two  or  more  adjacent  sections  of 
equal  size  and  number  of  turns.  As  in  the  series-bifilar  winding,  the  distributed  capacitance 
is  approximately  inversely  proportional  to  the  square  of  the  number  of  sections.  The 
turns  in  adjacent  sections  are  wound  in  opposite  directions  to  reduce  the  inductance, 
thereby  approximating  the  reversed  layers  of  a  reverse  layer  winding.  Generally  the  spool 
is  sectionalized  to  simplify  the  winding.  However,  if  the  wire  is  large  and  the  turns  cor- 
respondingly few,  the  same  effect  is  attained  in  a  bunch-type  winding  in  which  the  wire 
is  wound  in  successive  bunches  around  the  core. 

Reverse  Half-section  Winding.  Although  the  reversed  section  winding  gives  excellent 
results  for  capacitance  reduction,  the  magnetic  coupling  between  adjacent  sections  is  so 


COMPOSITION  CARBON  RESISTORS  3-11 

low  that  in  spite  of  reverse  turns  the  inductance  is  often  too  high.  This  is  especially  true 
for  values  ranging  from  1000  to  4000  ohms.  Although  above  this  upper  limit  high  residual 
inductance  still  obtains,  tolerable  phase  angle  results  because  of  the  increasing  effect  of 
higher  resistance  value  in  the  L  —  CRZ  relationship  of  eq.  (6) .  Reversing  by  half -sections 
has  the  effect  of  increasing  the  capacitive  and  decreasing  the  inductive  residuals.  This 
winding  is  somewhat  more  expensive  to  apply  than  the  reverse  section  type. 

Parallel  Opposing  (Ayrton-Perry)  Winding  (See  Fig.  14).  This  type  of  winding  consists 
of  two  inductive  windings  on  a  core  with  turns  equal  and  in  opposite  direction.  A  spaced 
layer  of  either  insulated  or  bare  wire  is  first  applied  in  one  direction.  The  second  wire  is 
wound  in  the  opposite  direction  between  the  turns  of  the  first  winding.  When  bare  wire 
is  used,  the  cross-overs  must  be  at  Tocommon 

exactly  diametrically  opposite  Sides  Of  terminal      Crossovers  (pomts  of  equal  potential  on  each  wire) 

the   core   so   that   contact   occurs   at  \ 

points  of  equal  potential  on  each  wire.     /^-^\  Y 

The  two  windings  are  connected  in 

parallel.     The  distributed  capacitance 

is  low,  and  the  opposing  currents  in 

the  two  wires  produce  the  minimum  "  N^  To  cornrn'OQ 

Of   magnetic    effect.      It    has   the    dis-  Two  wires  wound  on  core  in  smgje  layer  in        terminal 

advantage  of  requiring  four  times  as  ^^SH^^S'" tim 

much  wire  of  any  given  size  to  obtain  circumference 

the  same  value  produced  from  a  nor-      FlQ>  M>    ParaUel  Opposing  (Ayrton-Perry)  Winding 

mal  single-wire  winding. 

Mandrelated  Filament  Winding.  This  is  an  adaptation  of  the  continuous  winding  dis- 
cussed earlier,  in  which  a  single  layer  of  resistance  wire  is  wound  on  a  flexible  core.  The 
helical  filament  so  formed  can  then  be  handled  in  much  the  same  manner  as  ordinary 
resistance  wire.  A  number  of  such  filaments  have  been  developed  with  resistance  wire, 
dimensions,  and  turn  spacing  so  selected  that  for  any  straight  length  of  filament  the  in- 
ductive component  L/R  is  substantially  equal  to  the  capacitive  component  CR.  The 
effect  of  winding  on  a  form  and  terminating  the  filament  is  to  increase  the  capacitance 
slightly.  Because  of  the  excellent  frequency  characteristics  and  the  high  resistance  per 
linear  length  of  filament  (up  to  2400  ohms  per  inch  in  0.030-in.  diameter  of  insulated 
filament),  this  type  of  winding  is  widely  used  for  high  resistance  values  in  many  of  the 
designs  developed  for  use  in  the  communications  field. 

Miscellaneous  Windings.  Numerous  other  windings  have  been  developed  for  use  at 
high  frequency  that  serve  satisfactorily  but  are  not  in  general  commercial  use  except  in 
precision-type  measuring  apparatus  and  in  resistance  standards.  One  of  these  is  the 
"woven  type"  in  which  wire  is  woven  into  a  cloth  or  ribbon  pattern  giving  the  effect  of  a 
continuous  winding  on  a  card  of  infinitesimal  thickness.  Rather  intricate  patterns  are 
sometimes  used  to  obtain  low  phase  angle.  Constructions  such  as  "reversed  turn"  and 
"Curtis"  windings  require  handling  of  each  turn  individually  and  therefore  do  not  lend 
themselves  to  economic  manufacture  in  mass  production.  Occasionally  zero  phase  angle 
is  achieved  by  adding  the  necessary  amount  of  capacitance  or  inductance  to  a  completely 
wound  unit. 

3.  COMPOSITION  CARBON  RESISTORS 

FIXED  RESISTANCE  RESISTORS.  The  designation  "composition  resistor"  denotes 
a  type  of  resistor  that  has  very  wide  application  in  electronic  equipment  and  apparatus 
because  of  its  light  weight,  compactness,  wide  range  of  resistance  values  covered,  and  ease 
of  mounting.  It  is  primarily  used  in  circuits  where  drift  and  variation  in  resistance  value 
with  time,  temperature,  humidity,  and  applied  voltage  are  not  of  particular  significance. 
In  general,  the  resistive  element  in  a  composition  resistor  is  a  combination  of  finely  divided 
carbon  or  graphite,  a  non-conducting  inert  material  or  filler  such  as  talc,  with  synthetic 
resin  as  a  binder.  These  substances  are  proportioned  so  as  to  yield  the  proper  resistance 
value  in  the  finished  product. 

|l  Composition  resistors  are  available  in  insulated  and  non-insulated  types.  In  general, 
the  large  majority  of  composition  resistors  used  in  electronic  equipment  is  of  the  insulated 
type.  In  this  type  the  resistive  element  of  the  unit  is  surrounded  by  a  substantial  housing 
of  insulating  material,  such  as  mineral-filled  Bakelite,  so  that  there  is  no  possibility  of 
contact  with  the  element  other  than  through  the  wire  terminal  leads  of  the  resistor. 
The  usual  form  of  insulated  resistors  for  wattage  dissipations  of  2  watts  and  less  is  a 
cylindrical  body  provided  with  axial  terminals.  In  the  uninsulated  resistor,  a  cylin- 
drical rod  of  resistive  material  is  generally  equipped  with  radial  leads  and  the  unit  is 
painted. 


3-12 


RESISTORS,   INDUCTORS,   CAPACITORS 


SIZES  AND  RATINGS.     Typical  sizes  in  general  usage  of  fixed  insulated  axial  lead 
resistors  with  cylindrical  bodies  are  as  listed  in  Table  3. 

Table  3.     Sizes  of  Cylindrical  Insulated  Resistors 


Nominal 
Wattage 
Rating 
at  40°  C 

Maximum 
Body 
Length, 
in. 

Maximum 
Body 
Diameter, 
in. 

Lead 
Length, 
in. 

Min.  Lead 
Diameter, 
in. 

0.25 
0.50 
1.0 
2.0 

0.438 
0.438 
0.750 
0.750 

0.125 
0.156 
0.280 
0.344 

1  V2  ±  VS 
1  V2  ±  VS 
1  1/2  ±  1/8 

1  1/2  =fc  1/8 

0.028 
0.028 
0.032 
0.036 

The  resistors  listed  in  Table  3  are  available  in  resistance  values  ranging  ^  from  a  few 
ohms  to  22  megohms  on  a  manufacturer's  standard  basis,  with  the  qualification  that  the 

:.     Nominal  Resistance  Values 

(RMA  preferred  number  system) 


low  as  for  the  other  units. 
These  resistors  are  fur- 
nished in  accordance  with 
the  RMA  preferred  num- 
ber system  for  the  values 
and  tolerances  shown  in 
Table  4. 


jmber  of  zeros  or 
decimal  multiplier 


,       , 

I     2nd  )  significant 

'     1st  3 


figures 

FIG.  15.    Standard  Color  Cod- 
ing 

The  resistance  value  is 
indicated  by  a  color  code 
applied  to  the  resistor  as 
shown  in  Figs.  15  and  16 
and  Table  5.  The  exterior 
body  color  of  insulated  re- 
sistors may  be  any  color 
except  black.  It  is  recom- 
mended that  the  maxi- 
mum continuous  working 
voltage,  either  d-c  or  rms 
volts  for  the  0.5-watt  unit; 


Nominal  Resistance 

Available  in 
Tolerances 
±  per  cent 

ohms 

megohms 

10 

100 

,000 

10,000 

0.10,     .0,  10 

5,  10,  20 

n 

110 

,100 

11,000 

0.11,     .1,  11 

5 

12 

120 

,200 

12,000 

0.12,    .2,  12 

5,  10 

13 

130 

,300 

13,000 

0.13,     .3,  13 

5a 

15 

150 

,500 

15,000 

0.15,     .5,  15 

5,  10,  20 

16 

160 

,600 

16,000 

0.16,     .6,  16 

5 

18 

180 

,800 

18,000 

0.18,     .8,  18 

5,  10 

20 

200 

2,000 

20,000 

0.20,  2.0,  20 

5 

22 

220 

2,200 

22,000 

0.22,  2.2,  22 

5,  10,  20 

24 

240 

2,400 

24,000 

0.24,  2.4 

5 

27 

270 

2,700 

27,000 

0.27,  2.7 

5,  10 

30 

300 

3,000 

30,000 

0.30,  3.0 

5 

33 

330 

3,300 

33,000 

0.33,  3.3 

5,  10,  20 

36 

360 

3,600 

36,000 

0.36,  3.6 

5 

39 

390 

3,900 

39,000 

0.39,  3.9 

5,  10 

43 

430 

4,300 

43,000 

0.43,  4.3 

5 

47 

470 

4,700 

47,000 

0.47,  4.7 

5,  10,  20 

51 

510 

5,100 

51,000 

0.51,  5.1 

5 

56 

560 

5,600 

56,000 

0.56,5.6 

5,  10 

62 

620 

6,200 

62,000 

0.62,  6.2 

5 

68 

680 

6,800 

68,000 

0.68,  6.8 

5,  10,  20 

75 

750 

7,500 

75,000 

0.75,  7.5 

5 

82 

820 

8,200 

82,000 

0.82,  8.2 

5,  10 

91 

910 

9,100 

91,000 

0.91,  9.1 

5 

a-c,  should  not  exceed  250  volts  for  the  0.25-  watt  unit;  350 
and  500  volts  for  the  1-  and  2-watt  units. 


Table  5.     Color  Code 


Color 

Figure  or 
Number  of 
Zeros 

Decimal 
Multiplier 

Tolerance, 
per  cent 

Black  

0 

Brown 

1 

Red   ... 

2 

Orange  

3 

Yellow  

4 

Green 

5 

Blue 

6 

Violet  

7 

Gray  

8 

White 

9 

Gold  ..         .      . 

0.10 

±  5 

Silver  

0.01 

±10 

No  color  

±20 

Tolerance 


Significant 
f  igu  res 


FIG.  16.     Alternate  Color  Cod- 
ing for  Radial-lead  Resistors 


COMPOSITION  CARBON  RESISTOKS 


3-13 


PERFORMANCE  CHARACTERISTICS.  In  addition  to  obvious  requirements  on  the 
mechanical  properties  of  a  resistor,  such  as  ruggedness,  security  of  terminals,  legibility  of 
color  code,  and  its  d-c  resistance  value  and  tolerance  as  established  by  a  suitable  d-c 
resistance  measurement,  the  following  properties  are  of  interest  for  commercial  uses. 

Resistance-temperature  Characteristic.  Table  6  indicates  the  range  within  which  an 
insulated  composition  resistor  may  be  expected  to  vary  at  the  temperatures  indicated. 

Table  6.     Insulated  Composition  Resistor  Variations  Due  to  Temperature 


Nominal  Resistance  Value 

Maximum  Per  Cent  Change  in  Resistance  from  Value  at 
Ambient  Temperature  of  +25  °  C 

To  -15°C 
Ambient 

To  -  55°  C 

Ambient 

To  +65°C 
Ambient 

To  +105°C 
Ambient 

10  ohms  to  1000  ohms 

-0,     +3.5 
-0,     +5 
-0,     +6.5 
-0,  +10 

-0,  +  6.5 
-0,  +10 
-0,  +13 
-0,  +20 

±3 
±4.5 
±4.5 

±5 

±5 
±8.5 
±8.5 
±10 

1010  ohms  to  10  000  ohms  

1  0  1  00  ohms  to  0.  1  megohm  

Over  0.  1  megohm  

Voltage  Coefficient.  This  coefficient  relates  to  the  percentage  change  in  resistance  value 
per  unit  change  in  voltage  with  applied  voltage  as  distinguished  from  any  effects  caused 
by  heating  at  the  applied  voltage.  It  arises  from  a  change  in  the  conducting  properties 
of  the  resistive  material  as  the  applied  voltage  is  varied.  Voltage  coefficient  is  usually 
determined  for  resistors  of  1000  ohms  and  above  as  follows: 

Voltage  coefficient  (per  cent)  =  100  — ^-=r — -  X  -= =- 

where  E\  —  rated  continuous  working  voltage,  Bz  ~  0.1  rated  continuous  working  voltage, 
jRi  =  resistance  at  rated  continuous  working  voltage,  and  Rz  —  resistance  at  0.1  continuous 
working  voltage.  For  resistors  rated  at  1/4  and  1/2  watt,  the  voltage  coefficient  should 
not  exceed  0.035  per  cent  per  volt,  and  for  higher-wattage  resistors  it  should  not  exceed 
0.02  per  cent  per  volt. 

Humidity  Effects.  Test  data  indicate  that  molded-housing-type  insulated  resistors 
which  have  been  thoroughly  dried  and  then  subjected  to  a  condition  of  90  per  cent  relative 
humidity  at  an  ambient  of  30  deg  cent  for  200  hours  may  in  general  be  expected  to  stay 
within  a  limit  of  about  5  per  cent,  the  magnitude  of  the  change  depending  upon  resistance 
value.  The  product  of  certain  manufacturers  is  also  capable  of  withstanding  exposure  to 
100  per  cent  relative  humidity  at  an  ambient  temperature  of  66  deg  cent  for  250  hours 
without  changing  ha  resistance  value  by  more  than  10  per  cent. 

Noise.  Current  noise  arises  within  the  resistive  element  of  a  composition  resistor  pri- 
marily because  of  the  microphonic  nature  of  particle-to-particle  conduction  of  current 
in  the  structure.  In  some  resistors  noise  may  also 
originate  at  the  junction  between  the  lead  wire  and 
resistive  element  because  of  imperfect  contact. 

It  has  been  found  that  the  product  of  certain 
manufacturers  will  have  root-mean-square  values 
of  current  noise  with  rated  d-c  voltage  applied 
to  the  resistor  terminals  less  than  3  rms  micro- 
volts per  volt  for  1/2-watt  units  up  to  a  resistance 
value  of  1  megohm,  and  6  rms  microvolts  per  volt 
for  resistance  values  above  1  megohm.  For  1- 
and  2-watt  resistors,  the  noise  level  may  be  ex- 
pected not  to  exceed  1.2  rms  microvolts  per  volt. 

Load-life  Characteristic.  D-c  load  tests  con- 
ducted at  40  deg  cent  have  shown  such  extremely 
wide  variations  in  resistance  change  in  1000 
hours  that  no  general  statement  is  of  value. 
Units  have  been  found  to  age  either  positively 
or  negatively,  the  film-type  resistor  usually  show- 
ing an  increase  in  value  and  the  body  type  a 
decrease  in  value.  Hence  the  manufacturer 
should  be  consulted  for  specific  information  on 
the  performance  of  his  product.  However,  resistors  are  available  which  will  not  change 
more  than  10  per  cent  under  this  1000-hour  load  test. 

For  resistors  designed  to  carry  100  per  cent  of  rated  load  at  40  deg  cent  ambient,  it  is 
recommended  that  the  derating  curve  shown  in  Fig.  17  be  followed  to  prevent  undue  aging. 


UO 
100 


90 


'  70 


n-  30 
20 


10 


\ 


0    10  20  30  40   50  60  70  SO  90100110 
Ambient  temperature  in  degrees  centigrade 

FIG.  17.     Derating  Curve  for  High  Ambient 
Temperatures 


3-14 


RESISTORS,   INDUCTORS,   CAPACITORS 


It  will  be  noted,  for  example,  that  resistors  functioning  in  an  ambient  of  85  deg  cent  should 
be  operated  at  not  more  than  25  per  cent  of  rated  load.  Resistors  are  also  available  which 
may  be  operated  at  full  wattage  at  70  deg  cent  with  a  derating  to  zero  wattage  at  150  deg 
cent. 

Effect  of  Soldering.  Care  must  always  be  taken  to  prevent  the  quality  of  a  composition 
resistor  from  being  seriously  impaired  and  its  resistance  value  and  stability  affected  sig- 
nificantly by  excessive  heating  during  the  operation  of  soldering  its  terminal  leads  to 
apparatus  or  equipment  terminals.  The  leads  should  be  left  as  long  as  possible  and  prefer- 
ably not  less  than  3/8  in. 

Additional  Characteristics.  In  addition  to  the  properties  and  tests  described  above, 
other  characteristics  of  interest  in  military  work  and  of  possible  importance  in  certain 
commercial  application  may  be  mentioned:  voltage  breakdown  strength  of  the  insulation 
on  insulated-type  resistors;  high-altitude  flashover  voltage;  performance  after  salt-water- 
immersion  cycling;  effects  of  thermal  shock  under  temperature  cycling  in  air;  effect  of 
mechanical  vibration  on  mounted  resistors;  performance  under  other  types  of  load-life 
tests;  short-time  overload  performance;  and  ability  to  withstand  specific  mechanical  tests 
on  the  security  of  terminals.  Further  details  on  tests  and  expected  performance  may  be 
found  in  Specification  JAN-R-11. 

FREQUENCY  CHARACTERISTIC.  In  general  the  inductance  associated  with  com- 
position resistors  is  sufficiently  small  to  be  disregarded.  At  low  frequencies  the  value  of 
resistance  R  is  the  same  as  the  d-c  resistance,  but  as  the  operating  frequency  is  raised  the 
value  of  R  starts  to  decrease  and  may  reach  a  value  which  is  only  a  few  per  cent  of  the  d-c 
value.  The  frequency  at  which  R  begins  to  show  a  significant  decrease  in  value  depends 
upon  its  d-c  value;  that  is,  the  greater  the  d-c  value  of  a  resistor,  the  lower  the  frequency 
at  which  departure  from  the  d-c  value  is  observed.  Incidentally,  parallel  capacitance  C 
shows  a  similar  decline  in  value,  but  the  amount  of  change  is  significantly  less  than  the 
reduction  in  resistance. 

Figure  18  shows  the  variation  in  resistance  with  frequency  of  two  different  1-megohm 
insulated  resistors  of  i/a-watt  rating.  These  resistors  are  practically  identical  in  physical 

size,  but  one  has  a  resistive 
element  in  the  form  of  a 
composition  film  on  the  sur- 
face of  a  glass  tube  and  the 
other  has  a  resistive  element 
of  the  body  type.  Resistors 
that  have  been  found  to 
show  the  least  decrease  with 
frequency  have  a  very  thin 
uniform  resistive  film  on  a 
rod  or  tube  of  low-loss  di- 
electric material  in  which 
the  ratio  of  length  of  ele- 
ment to  its  diameter  is  large 
and  in  which  no  exterior 
covering  or  coating  material 
is  in  contact  with  the  sur- 
face of  the  element. 

It  has  been  observed  that,  in  a  given  specific  type  of  resistor,  the  ratio  of  parallel  a-c 
resistance  to  the  d-c  resistance  is  approximately  constant  for  a  fixed  value  of  the  product 
of  operating  frequency  and  d-c  resistance  value.  That  is  to  say  the  ratio  is  approximately 
the  same  for  a  10-megohm  unit  operating  at  0.1  me,  a  1-megohm  unit  at  1  me,  and  a  0.1- 
megohm  unit  at  10  me.  In  each  instance  the  product  of  megohms  and  megacycles  is 
unity.  Applying  this  example  to  the  curve  given  for  the  body-type  resistor  in  Fig.  18, 
and  noting  that  at  1  me  the  1-megohm  resistor  has  53  per  cent  of  its  d-c  resistance,  a 
resistor  having  a  value  of  10  megohms  (d-c)  and  being  of  the  same  construction  as  the 
1-megohm  unit  for  which  the  curve  is  drawn  would  have  a  parallel  resistance  of  around 
5.3  megohms  at  0.1  me.  Also  a  0.1-megohm  unit  (d-c)  at  10  me  would  have  a  value  of 
about  53,000  ohms  parallel  resistance.  It  is  very  important  to  observe  that  this  relation- 
ship is  only  approximate  and  also  must  be  established  for  each  specific  type  of  resistor.* 
This  is  evident  from  the  fact  that,  for  a  product  of  unity  for  the  film  type  of  resistor  shown, 
the  parallel  resistance  is  about  80  per  cent  of  the  d-c  value,  as  compared  to  53  per  cent  for 
the  body-type  unit. 

Table  7  indicates  the  approximate  frequency  at  which  a  1-megohm  resistor  will  have  a 
parallel  resistance  of  0.5  megohm  for  0.5-,  1-  and  2-watt  resistors  within  the  dimensional 
limits  stated  in  Table  3.  With  the  product  relationship  discussed  above,  the  data  in  Table  7 


J  0.9 
§0.8 

=  °'7 
tt*0.6 

Jo.4 
-£  0.3 
Jo.2 

<£      0 

••^^ 

•*-: 

--  — 

K 

c* 

ominal  rating,  %  watt 
ze,  see  table  1 
c  resistance,  1  megpl* 

m 

Ns 

*•-. 

>s 

X, 

^ 

^ 

>v 

s»s> 

^^ 

X" 

% 

"\ 

Vi 

4^ 

X. 

\ 

v^, 

*?°n 

^ 

)      2     4  6  10          40     100        400 
Kilocycles  per  second 

•       2  34  6    10   20   40    100 
Megacycles  per  second 

Frequency 

FIG.  18.     Variation  of  Parallel  Resistance  R  with  Frequency  for  1- 
megohm  Molded  Insulated  Housing  Type  of  Resistor 


DEPOSITED-CARBON  BESISTORS 


3-15 


may  be  used  to  estimate  the  frequencies  at  which  other  resistors  of  the  same  size  and  type 
will  drop  to  values  of  parallel  resistance  equal  to  half  of  the  d-c  resistance. 

The  parallel  capacitance  C  associated  with  these  resistors  is  also  a  function  of  resistor 
construction,  resistance  value,  and  frequency.  Considerable  variation  in  capacitance 
value  for  different  samples  of  the  same  type  has  been  noted,  but  the  value  appears  to  be 
in  the  order  of  1  to  3  ;Ujuf,  falling  with  frequency  to  around  0.5  to  1  Mjuf. 

HIGH-FREQUENCY  AND  HIGH-VOLTAGE  TYPES.  Another  line  of  composition- 
type  resistors  is  available  which  is  designed  for  high-voltage  and  high-frequency  applica- 
tions. These  have  various  wattage  ratings 

Table 


7.     Resistance-frequency   Character- 
istics of  Specific  Resistors 


1  -megohm  Resistor 
Nominal  Wattage 
Rating 

Frequency  in  Megacycles 
for  R  of  0.5  Megohm 

Film-type 
Element 

Body-type 
Element 

0.5 
1.0 
2.0 

20 
3.0 

1.2 
0.8 
0.4 

from  less  than  1  watt  to  about  100  watts. 

In  general  the  high-frequency  type  con- 
sists of  a  continuous  film  of  resistive  element 
on  the  surface  of  a  ceramic  tube.  In  size,  as 
typified  by  the  product  of  one  manufac- 
turer, these  resistors  range  from  units  about 
1/8  in.  in  diameter  and  lli  in.  long  with  wire 
lead  axial  terminals  to  units  about  2  in.  in 
diameter  and  20  in.  long  overall,  including 
ferrule  terminals.  Resistance  values  of  a  few 
ohms  to  several  megohms  are  available,  de- 
pending upon  the  physical  size  of  the  resistor. 

High-voltage  composition-type  resistors  are  of  similar  construction  to  that  described 
for  the  high-frequency  type,  except  that  the  resistive  element  is  usually  in  the  form  of  a 
spiral  or  ribbon  to  provide  a  long  conducting  path.  They  are  available  in  sizes  ranging 
from  units  5/l6  in.  in  diameter  and  about  2  in.  long  rated  at  2  watts  and  7500  volts  maxi- 
mum, to  units  2  in.  in  diameter  and  about  20  in.  long  rated  at  150  watts  and  100,000  volts 
maximum.  Resistance  values  obtainable  range  from  a  few  thousand  ohms  to  a  million 
megohms. 

4.  DEPOSITED-CARBON  RESISTORS 

GENERAL  DESCRIPTION.  "  Deposited-carbon  resistor"  denotes  a  kind  of  resistor 
in  which  the  resistive  element  is  a  film  of  carbon  deposited  on  the  surface  of  a  suitable 
ceramic  core  by  the  thermal  decomposition  of  gaseous  hydrocarbons  at  high  temperatures. 
This  film  is  extremely  thin  and  by  proper  control  of  the  coating  process  may  be  varied  in 
thickness  within  the  range  from  1  X  10  ~4  to  5  X  10  ~8  in.  The  resistance  of  such  films 
ranges  from  about  5  ohms  per  unit  square  to  about  10,000  ohms  per  unit  square.  "Ohms 
per  unit  square"  denotes  the  resistance  as  measured  between  opposite  edges  of  a  square 
of  resistance  film  of  the  thickness  indicated. 

Electrical  connection  is  made  to  the  carbon  film  by  applying  low-resistance  electrodes 
of  either  graphitic  or  special  metallic  paint.  These  electrodes  are  cured  by  suitable  heat 
treatment,  and  then  metal  caps  with  integral  lead  wires  are  forced  over  them.  Since  the 
carbon  film  is  sensitive  to  abrasion  and  also  needs  protection  from  contamination,  the 
resistance  structure  is  coated  with  a  suitable  baking  varnish  or  housed  in  an  enclosure. 
Resistors  having  resistance  values  up  to  a  few  thousand  ohms  are  formed  of  uniform  films. 
High-ohmage  units  are  obtained  by  cutting  a  helical  groove  through  the  carbon  film  to 
form  a  ribbon  of  film  wound  around  the  core  between  the  end  electrodes.  The  surface 
perfection  of  the  ceramic  core  and  its  other  physical  and  chemical  properties  have  a  marked 
effect  on  the  electrical  characteristics  of  the  resistor. 

USAGE.  Deposited-carbon  resistors  provide  exceptional  resistance  stability  and  com- 
pactness in  high  values  of  resistance.  Also,  because  of  low  residual  inductance,  the  power- 
type  varieties  are  of  value  as  load  resistors  in  testing  high-frequency  equipment.  Special 
shapes  of  deposit ed-carbon  resistors  in  the  form  of  suitably  terminated  small  rods  and 
disks  are  available  for  assembly  in  coaxial-type  attenuator  units  for  use  in  making  accurate 
attenuation  measurements  up  to  frequencies  of  several  hundred  megacycles. 

SIZES  AND  RATINGS.  Typical  figures  for  the  product  of  one  manufacturer  are  given 
in  Table  8.  These  resistors  have  cylindrical  bodies.  The  resistors  with  the  shell  enclosures 
have  axial  leads  and  are  intended  for  general-purpose  use.  The  glass-enclosed  units  are 
hermetically  sealed  with  an  inert  gas  in  the  enclosure  and  are  provided  with  ferrule  end 
caps  as  terminals;  they  are  intended  for  applications  in  which  high  stability  in  resistance 
value  or  high  levels  of  power  dissipation  are  involved. 

PERFORMANCE  CHARACTERISTICS.  The  general-purpose  resistors  and  the 
glass-enclosed  types  for  high-stability  application  are  available  in  tolerances  as  close 
as  ±1  per  cent.  The  large  power  dissipating  units  are  available  in  tolerances  of  ±5 
per  cent. 


3-16 


KESISTORS,   INDUCTORS;   CAPACITORS 


Table  8.    Ratings  and  Sizes  of  Deposited-carbon  Resistors 


Rating 
Norn, 
at  30°  C 

Wattage, 
Maximum  * 

Approximate  Overall 
Size,  in. 

Resistance  Range, 
ohms 

Short- 
period 
Peak 
Voltage  f 

Protective 
Enclosure 

Diameter 

Length 

Minimum 

Maximum 

0.15 
0.5 
1.0 
1.0 
2.0 

0.5 
1.5 
3.0 
10 
20 
60 
300 
600 

H/64 
H/32 
H/32 
7/16 
7/16 
5/8 
11/4 
U/4 

Wl6 
1 

2Vl6 
21/4 
3V4 
411/16 
83/4 
143/4 

1 
200 
200 
200 
200 
20 
20 
40 

5X  106 
106 
5  X  107 

107 

1.5  X  10? 

10? 

5X  106 

107 

5,000 
8,000 
15,000 
2,000 
6,000 
10,000 
20,000 
40,000 

Shell 
Shell 
Shell 
Glass 
Glass 
Glass 
Glass 
Glass 

*  At  maximum  wattage,  resistance  may  differ  from  30  deg  cent  value  by  decreases  of  10  to  15  per  cent, 
t  Not  to  exceed  that  required  for  maxim  urn  power  rating. 

Temperature  Coefficient.  The  temperature  coefficient  of  the  carbon  film  depends  upon 
its  thickness  and  ranges  from  about  — 180  parts  per  million  per  degree  centigrade  for  very 
heavy  coatings  to  about  —  500  parts  per  million  per  degree  centigrade  for  light  coatings. 
Furthermore,  the  application  of  protective  lacquer  to  the  film  may  modify  its  temperature 
coefficient  by  virtue  of  mechanical  effects  on  the  film.  The  physical  properties  of  the 
ceramic  base  also  affect  this  coefficient.  Hence  the  temperature  coefficients  of  these 
resistors  range  from  about  —0.02  to  possibly  —0.10  per  cent  per  degree  centigrade,  depend- 
ing upon  resistance  value  and  constructional  features.  The  resistance-temperature  curve 
is  approximately  linear  over  the  temperature  range  of  —40  deg  cent  to  +60  deg  cent. 

Voltage  coefficient  is  in  general  negligible  for  deposited-carbon  units.  Occasionally  an 
individual  resistor  may  show  a  slight  resistance  variation  with  voltage  but  probably  not 
more  than  0.002  per  cent  per  volt. 

Humidity  effects  depend  upon  the  structure  of  the  resistor  and  in  hermetically  sealed 
units  become  a  matter  of  leakage  across  the  surface  of  the  housing.  The  following  figures 
are  indicative  for  general-purpose  units  with  shell  enclosures.  After  exposure  to  a  condition 
of  90  per  cent  relative  humidity  at  an  ambient  of  30  deg  cent  for  200  hours,  the  maximum 
change  in  resistance  may  be  expected  to  be  less  than  1  per  cent,  with  an  average  change 
of  less  than  0.5  per  cent. 

Noise.  At  low  levels  of  voltage,  deposited-carbon  resistors  exhibit  pure  thermal  noise, 
but  as  the  voltage  is  raised  other  electrical  noise  may  be  observed.  This  may  be  some  form 
of  contact  noise  due  to  imperfections  or  loose  contacts  in  the  carbon  film.  However,  at 
rated  load  the  noise  level,  excluding  thermal  noise,  may  be  expected  not  to  exceed  0.25 
rms  microvolt  per  volt. 

Load-life  Characteristics.  General-purpose-type  deposited-carbon  units  when  operated 
at  their  normal  wattage  rating  may  be  expected  to  show  average  changes  in  resistance 
value  of  not  more  than  1  per  cent  after  2000  hours.  Occasional  units  may  age  as  much  as 
1.5  per  cent.  Hermetically  sealed  resistors  operated  at  high  power  levels  may  show  more 
rapid  aging.  For  example,  the  7/i6  X  3  1/4  unit  in  Table  8  may  change  in  resistance  value 
up  to  3  per  cent  after  3000  hours  of  operation  at  10  watts.  Load  aging  may  cause  either  a 
positive  or  negative  change  in  resistance  but  generally  results  in  a  positive  change.  In 
using  power-type  resistors,  care  must  be  taken  to  insure  that  the  voltage  gradient  within 
the  resistor  is  not  sufficient  to  give  rise  to  corona  or  cause  flashover  between  adj  acent  turns 
of  a  spiraled  element  and  thus  damage  the  unit. 

Like  composition  resistors,  when  deposited-carbon  units  are  operated  in  high  ambient 
temperatures  they  should  be  derated.  For  types  which  are  not  hermetically  sealed,  it  is 
recommended  that  the  maximum  operating  surface  temperature  of  the  carbon  film  not 
exceed  120  deg  cent.  Since  their  normal  rating  is  established  at  30  deg  cent,  the  wattage 
ratings  for  these  units  should  be  decreased  by  about  1  per  cent  for  each  degree  centigrade 
that  the  ambient  exceeds  30  deg  cent.  For  glass-sealed\mits,  which  under  maximum  power 
ratings  may  operate  at  surface  temperatures  up  to  450  deg  cent,  the  ratings  can  be  re- 
garded as  independent  of  ambient  temperature  up  to  about  80  deg  cent.  The  manu- 
facturer should  be  consulted  for  further  derating  information  for  the  specific  conditions 
of  application.  The  ratings  of  power  units  can  be  increased  several  fold  by  forced  air  cool- 
ing, and,  for  a-c  application,  resistors  are  available  which  may  be  liquid  cooled  through 
direct  contact  of  the  coolant  with  the  film.  Anodic  oxidation  of  the  film  precludes  the 
use  of  water  cooling  on  d-c  applications. 

The  effect  of  soldering  is  negligible  on  lead-type  units,  provided,  of  course,  that  the  re- 
sistor is  not  damaged  by  direct  contact  with  the  soldering  tool. 


POTENTIOMETERS  AND  RHEOSTATS         3-17 

No-load  Aging.  Under  conditions  of  no-load  or  shelf-aging,  deposited-carbon  resistors 
of  the  general-purpose  type  may  be  expected  to  drift  in  value  not  more  than  0.1  to  0.2 
per  cent  per  year.  Hermetically  sealed  units  appear  to  have  a  stability  in  resistance  value 
of  the  order  of  0.005  per  cent  per  year,  which  is  about  the  limit  of  error  in  measurements 
extending  over  such  a  time  period.  In  fact  the  stability  of  high-quality  hermetically  sealed 
carbon  film  resistors,  particularly  in  the  megohm  region,  appears  to  be  at  least  as  good  as, 
if  not  better  than,  that  of  equivalent  resistance  wire-wound  units. 

FREQUENCY  CHARACTERISTIC.  Deposited-carbon  resistors  exhibit  a  decrease  in 
parallel  resistance  with  frequency.  However,  the  rate  of  decrease  does  not  appear  to  be 
as  rapid  as  in  the  insulated-type  composition  resistor,  and  it  is  less  for  the  glass-enclosed 
type  than  for  the  varnished  or  shell-protected  unit  in  which  the  protecting  material  is  in 
contact  with  or  very  close  to  the  carbon  film.  A  1-watt  varnish-coated  1-megohm  unit, 
9/32  in.  in  diameter  and  2  1/16  in.  long,  drops  to  0.5  megohm  parallel  resistance  at  about 
12  me. 

Whereas  the  inductance  of  an  unspiraled  film  is  negligible,  the  inductance  of  a  spiraled 
unit  may  be  appreciable.  For  example,  the  inductance  of  a  1-megohm  resistor  of  the 
7/i e  in.  X  3  1/4  in.  size  in  Table  8  is  about  1.1  microhenries.  Though  spiraling  increases 
inductance,  the  ratio  of  inductance  to  resistance  remains  essentially  unchanged.  Since 
the  effects  of  distributed  capacitance  are  changed  only  slightly  by  spiraling,  the  alteration 
in  high-frequency  behavior  of  a  resistor  which  is  spiraled  to  a  high  resistance  value  is 
largely  that  associated  with  the  resistance  increase  alone. 

5.  METAL  FILM  RESISTORS 

These  resistors  are  formed  by  coating  a  base  of  suitable  material  with  a  very  thin  film 
of  metal  or  metallic  alloy.  This  film  may  be  applied  to  the  base  by  cathode  sputtering  or 
metal  evaporation  processes,  and  by  chemical  methods.  It  is  quite  thin,  being  of  the  order 
of  10 ~7  in.  thick.  The  material  used  for  the  base  may  be  ceramic,  glass,  or  an  organic  body 
such  as  Bakelite. 

In  one  construction,  a  thin  film  of  palladium  is  deposited  by  a  chemical  process  on  a 
ceramic-core  tube.  The  ends  of  the  film  are  coated  with  silver  or  some  other  metal  to 
form  electrodes  for  the  attachment  by  soldering  of  radial  lead  wires.  The  film  is  covered 
by  a  form  of  vitreous  covering  to  afford  protection  against  oxidation  and  corrosion  as  well 
as  against  mechanical  damage.  High  values  of  resistance  are  obtained  by  cutting  a  helix 
through  the  metal  film  into  the  core,  in  the  same  fashion  as  deposited-carbon  film  resistors 
are  helixed.  For  further  protection  the  whole  unit,  including  the  portion  of  the  lead  wires 
in  contact  with  the  resistor  body,  is  lacquered.  Resistors  made  in  this  way  are  available 
over  a  range  of  resistance  values  in  several  sizes  corresponding  to  different  wattage  ratings. 
They  possess  good  electrical  stability  under  aging  and  load  conditions,  and  they  have  good 
heat-dissipating  properties.  As  determined  from  observations  on  a  limited  number  of 
samples  with  a  nominal  rating  of  1  watt,  aging  under  d-c  rated  load  is  predominately 
positive  as  regards  resistance  value  and  may  be  of  the  order  of  1  to  5  per  cent  after  1000 
hours,  depending  upon  resistance  value.  The  temperature  coefficient  is  negative  and 
varies  to  some  extent  with  resistance  value.  It  also  shows  considerable  departure  from 
linearity  over  the  temperature  range  of  —40  deg  cent  to  +80  deg  cent.  Referred  to  resist- 
ance value  at  20  deg  cent,  resistors  increase  in  value  by  about  5.5  per  cent  at  —  40  deg  cent, 
and  at  80  deg  cent  they  decrease  in  value  by  about  3  per  cent. 

In  another  construction  of  metal  film  resistor,  a  resistance  alloy  such  as  Nichrome  is 
evaporated  to  form  a  very  thin  film  on  the  surface  of  a  glass  tube  or  rod.  A  layer  of  pro- 
tective material  is  placed  over  this  film,  and  lead  wires  are  attached  in  the  same  general 
manner  as  described  above.  An  advantage  of  this  type  of  construction  is  that  the  low 
temperature  coefficient  of  the  resistance  alloy  is  retained  in  the  resistor. 

6.  POTENTIOMETERS  AND  RHEOSTATS  * 

Potentiometers  and  rheostats  in  general  use  in  the  communications  and  electronics 
fields  may  be  classified  as  wire-wound  and  composition  types.  Those  most  extensively 
used  are  the  continuously  adjustable  type,  where  a  movable  contact  traverses  a  resistance- 
wire  winding  or  a  composition  resistance  element  in  small  increments  of  their  lengths.  In 
most  instances  the  movable  contact  is  controlled  by  a  rotatable  shaft,  although  in  some 
the  resistance  element  is  rotated.  It  is  standard  practice  to  have  terminals  for  both  ends 

*  Article  6  was  contributed  by  A.  H.  Vblz. 


3-18 


RESISTORS,  INDUCTORS,   CAPACITORS 


100 


"40  50    60    70  80    90  100 
Ambient  temperature  In 
degrees  centigrade 

FIG.     19.     Power     Derating 
Curve  for  Continuous  Duty 


of  the  resistance  element  in  addition  to  a  terminal  for  the  movable  contact  so  that  they 
may  be  wired  either  as  potentiometers  or  rheostats. 

Most  of  the  potentiometers  and  rheostats  hereinafter  described  may  be  obtained  in 
tandem  arrangement  either  with  a  common  shaft  which  adjusts  each  unit  simultaneously 

or  with  concentric  shafts  which  permit  independent  adjustment  of  the  tandem  units. 
WIRE-WOUND  POTENTIOMETERS  AND  RHEOSTATS.    Low-operating-tempera- 

ture potentiometers  find  application  as  voltage-  and  current-adjusting  devices  in  low- 
energy  circuits.  The  power  ratings  are  based  on  operation  in 
free  still  air  at  an  ambient  temperature  of  40  deg  cent  with  a 
maximum  temperature  rise  of  60  deg  cent.  A  power  derating 
curve  is  shown  in  Fig.  19.  When  potentiometers  are  enclosed 
and  in  close  proximity  to  other  components  it  is  considered 
good  practice  to  limit  the  power  dissipation  to  about  one-half 
the  rating. 

The  types  that  find  the  widest  application  are  ^  the  small 
circular  potentiometers  that  range  from  about  1  1U  in.  to  2  in. 
in  diameter  and  are  rated  from  about  2  to  4  watts.  -  In  this 
type  bare  wire  is  space  wound  on  a  flat  strip  of  insulating  mate- 
rial, usually  a  laminated  phenolic,  which  is  then  formed  into 
a  circular  shape  and  set  into  a  housing  or  case.  '  The  contact 
shoe  or  brush  is  made  of  a  base  metal  or  an  alloy  of  base  metals, 
supported  on  a  spring  member.  The  force  exerted  on  the 
resistance  wire  by  the  base-metal  contacts  usually  ranges  from 
about  100  to  200  grams,  sufficient  to  keep  the  contact  resist- 
ance below  1  ohm,  providing  a  satisfactory  level  of  contact 
noise  for  most  applications  and  a  useful  life  of  25,000  to 
100,000  cycles  of  operation.  Individual  potentiometer  designs 
may  show  changes  in  resistance  from  1  to  10  per  cent  over 

the  ranges  of  temperature  and  humidity  usually  encountered.  * 

In  general  it  is  desirable  to  use  no  smaller  than  1.75  mil  wire  (usually  similar  to  Ni- 

chrome),  resulting  in  maximum  resistance  of  about  50,000  ohms  for  the  2-in.  -diameter 

size  and  10,000  ohms  for  the  1  1/4-in.  size.    Standard  tolerances  on  resistance  are  ±10  per 

cent.    Closer  tolerances  and  higher  resistance  values  can  be  obtained  on  a  custom  basis-. 

Finer  wire  is  more  susceptible  to  wear  and  electrolytic  action. 

Various  resistance-rotation  characteristics  are  available,  the  most  extensively  used 

being  the  linear  type  wherein  the  rate  of  change  of  resistance  between  the  contact  terminal 

and  one  of  the  end  terminals  is  approximately 

constant  with  angular  rotation.    In  these  small 

types  the  degree  of  linearity  is  usually  of  the 

order  of  ±5  per  cent  of  total  resistance.    This 

type  of  characteristic  is  illustrated  by  curve  A 

of  Fig.  20. 

For  special  applications,  non-linear  or  ta- 

pered potentiometers  can  be  "obtained.     This 

non-linear  characteristic  is  effected  by  chang- 

ing the  pitch  of  the  winding  at  certain  points 

in  the  range  of  rotation  or  by  changing  wire 

size  or  by  a  combination  of  both.    Curve  B  of 

Fig.  20  illustrates  a  clockwise  and  non-linear 

characteristic  in  which  there  are  two  sections 

differing  in  their  rates  of  change  of  resistance. 

As  a  high  percentage  of  the  total  resistance 

may  be  concentrated  in  a  small  portion  of  the 

resistance   element,    it   is  recommended  that 

non-linear  controls  be  rated  at  about  half  of 

the  rated  power  of  linear  controls  of  the  same 

designs. 


80 


60 


10 


lockwise  rotation' 


Cur 


I 


Left 
terminal 


20          40   50    60  80 

Per  cent  effective  rotation 


| 


100 


-TotaJ  mechanical  rotation  — 
FIG.  20.    Clockwise  Tapers 


For  continuous  operation  of  linear  types,  the  maximum  current  through  the  entire  re- 
sistance element  or  through  any  portion  thereof  should  not  exceed  the  value  given  by  the 
following  equation: 


where  W  is  the  wattage  rating,  R  is  the  total  resistance,  and  I  is  the  maximum  permissible 
current.    The  maximum  current  for  each  section  of  a  non-linear  control  should  be  deter- 


POTENTIOMETERS  AND  RHEOSTATS  3-19 

mined  from  the  above  expression  except  that  R,  instead  of  being  the  actual  total  resist- 
ance, should  be  that  value  which  would  obtain  if  the  entire  resistance  element  were  wound 
with  the  same  size  wire  as  the  particular  section. 

There  is  another  group  of  circular,  wire-wound,  low-operating-temperature  potentiom- 
eters which  are  approximately  3  in.  in  diameter  and  range  from  about  1  1/4  in.  to  2  &/8  in. 
in  depth,  and  in  power  rating  from  8  to  15  watts.  The  winding  is  usually  clamped  around 
a  molded  phenolic  base.  This  class  of  potentiometer  is  usually  provided  with  a  winding 
of  a  higher  degree  of  uniformity  than  the  smaller  types,  owing  to  the  use  of  higher-precision 
winding  methods.  As  a  result  linearly  wound  potentiometers  are  obtainable  with  linearity 
ranging  from  1  to  0.3  per  cent.  Non-linear  types  are  usually  produced  by  shaping  the  card 
or  strip  on  which  the  wire  is  wound,  which  is  possible  because  of  the  greater  depth  of  the 
winding.  The  amount  of  taper  is  limited  by  the  maximum  slope  (angle  between  winding 
axis  and  shaped  side  of  card)  that  can  be  wound  without  having  the  winding  collapse 
which  for  close  winding  is  about  40  deg.  The  deepest  linear  unit  of  this  group  can  be  wound 
to  about  200,000  ohms  without  resorting  to  Nichrome  wire  finer  than  1.75  mil  in  diam- 
eter. These  types  are  readily  obtainable  with  low-temperature-coenicient  wire  such  as 
Advance,  which,  however,  will  wear  more  rapidly  than  Nichrome  since  it  is  softer.  An 
idea  of  the  performance  capabilities  of  potentiometers  of  these  two  groups  can  be  obtained 
by  referring  to  the  Joint  Army-Navy  Specification  JAN-R-19. 

Very  special  high-precision,  low-operating-temperature  wire-wound  potentiometers 
have  been  developed  for  military  applications.  Some  of  them  have  winding  cards  shaped 
to  provide  special  resistance-rotation  characteristics  to  an  accuracy  represented  by  two 
turns  of  the  winding  at  any  point  in  the  range  of  rotation.  Some  types  have  been  provided 
with  closed  windings  for  continuous  rotation;  in  one  such  type  an  input  d-c  voltage  is 
applied  through  two  fixed  taps  180  deg  apart  and  the  output  voltage  is  obtained  from  two 
rotating  brushes  diametrically  opposed  to  each  other.  If  the  winding  is  linear,  varying 
the  position  of  the  brushes  varies  the  output  voltage  in  accordance  with  a  linear  sawtooth 
wave. 

Many  of  these  types  have  been  provided  with  precious-metal  contacts  developed  spe- 
cifically to  obtain  low  contact  resistance  with  low  contact  pressure.  Contact  resistance 
of  a  few  hundredths  of  an  ohm  with  contact  pressures  of  about  50  to  80  grams  has  been 
obtained,  and  a  life  of  the  order  of  a  million  operations  has  been  realized.  One  type  of 
contact  alloy  widely  used  is  Paliney  No.  7,  which  consists  of  platinum,  palladium,  gold, 
silver,  copper,  and  zinc. 

In  another  type  of  low-operating-temperature  wire-wound  potentiometer,  known  as  a 
multiturn  potentiometer,  the  resistance  wire  is  wound  on  an  insulated  metal  mandrel 
about  1/s  in.  in  diameter  which  is  then  formed  into  a  helix.  The  diameter  and  number  of 
turns  of  the  helix  vary,  and  potentiometers  have  been  produced  commercially  with  2,  10, 
15,  25,  and  40  turns  and  with  overall  diameters  of  approximately  2,  3,  4,  and  6  in.  The 
contact  is  arranged  to  follow  the  path  of  the  helix.  With  this  type  very  fine  adjustments 
are  possible  because  of  the  comparatively  long  winding.  By  maintaining  close  tolerance 
on  the  diameter  of  the  mandrel  and  on  the  resistivity  and  diameter  of  the  wire,  linearity 
of  the  order  of  ±0.1  per  cent  or  better  can  be  provided. 

The  straight  winding  type  potentiometers  heretofore  described  are  not  very  suitable  for 
high-frequency  applications.  At  frequencies  above  the  audio  range  the  straight  winding 
controls  are  affected  by  distributed  capacitance  and  inductance.  For  higher-frequency 
applications  the  characteristics  of  the  individual  control  should  be  investigated. 

POWER-TYPE  RHEOSTATS.  Toroidal  Winding  Type.  This  type  is  the  most  ex- 
tensively used  for  applications  from  25  to  1000  watts  in  the  communications  and  electronics 
fields.  Adjustable  resistors  of  this  type  generally  consist  of  a  toroidal  ceramic  form  which 
is  wound  with  either  round  or  ribbon-type  wire  over  an  arc  of  approximately  300  deg. 
The  wound  form  is  then  placed  in  a  suitable  ceramic  base,  and  the  entire  unit,  except  for 
the  contacting  surface  of  the  wire,  is  given  a  coating  of  vitreous  enamel  under  high  tem- 
perature. This  form  of  construction  permits  a  high  wattage  rating  in  a  relatively  small 
volume  of  space. 

Rheostats  of  this  type  are  available  in  sizes  ranging  from  approximately  1  Va  in.  to  12 
in.  in  diameter  and  in  rated  wattage  from  25  to  1000  watts.  They  are  wound  to  resistance 
values  from  a  fraction  of  an  ohm  up  to  10,000  ohms.  The  power  rating  of  rheostats  is 
based  on  temperature  rise  in  free  still  air.  For  those  rated  at  100  watts  or  less,  the  tempera- 
ture rise  is  limited  to  300  deg  cent;  for  those  rated  above  100  watts  the  permissible  tem- 
perature rise  is  350  deg  cent. 

These  rheostats  are  available  with  either  linear  or  tapered  windings.  Specific  uses  for 
tapered  rheostats  are:  (1)  to  provide  a  more  uniform  degree  of  control  for  all  positions  of 
the  contact,  (2)  to  make  possible  the  use  of  a  smaller  control,  (3)  to  make  it  possible  to 
wind  a  higher  resistance  on  a  small  control  for  specific  applications,  (4)  to  provide  a  par- 


3-20 


RESISTORS,   INDUCTORS,    CAPACITORS 


§> 


ticular  controlled  effect.  An  example  of  this  last  might  be  to  provide  a  linear  relationship 
between  control  setting  and  motor  speed  in  the  case  of  motor  speed  controls,  or  to  give 
linear  control  of  light  output  from  a  lamp.  Figure  21  shows  how  the  current  varies  (in 
three  typical  rheostats)  with  per  cent  rotation  of  the  contact. 

For  special  applications,  controls  can  be  obtained  with  continuous  360-deg  windings, 
built-in  toggle  switches,  or  off  positions  at  either  end  of  the  rotation. 

Metal  Type.  Another  power  type  of  rheostat  utilizes  mostly  metal  in  its  construction. 
The  wire  or  ribbon  is  wound  on  a  strip  of  aluminum  with  asbestos  as  insulation  between 
the  wire  and  aluminum  strip.  The  winding  is  formed  into  a  circular  shape  and  is  assembled 

in  a  die-cast  aluminum  base  with  mica  sepa- 
rating the  winding  from  the  base.  Owing  to 
the  close  proximity  of  the  winding  to  the 
aluminum  parts,  the  heat  is  carried  away 
from  the  winding  more  rapidly  than  in  the 
ceramic  types.  As  a  result,  for  the  same 
wattage  dissipation  the  temperature  rise  is 
somewhat  lower. 

Standardization  requirements  and  per- 
formance capabilities  of  power-type  rheostats 
for  use  in  military  applications  are  con- 
tained in  Joint  Army-Navy  Specification 
JAN-R-22. 

Tubular  Slide-wire  Type.  The  tubular 
slide-wire  type  of  rheostat  is  used  extensively 
for  general  laboratory  applications,  particu- 
larly for  precision  measurements.  They  are 
not  generally  used  in  commercial  applica- 
tions, inasmuch  as  they  require  considerably 
more  mounting  space  for  equivalent  wattage 
ratings  and  are  not  as  convenient  as  the 
toroidal  type. 


V 

(c 

) 

75 

\ 

X 

3  sectio 

n  taper 

56.2 

\ 

\v 

\ 

•s. 

( 

B) 

i 

\ 

X 

2  sect) 

on  taper 
/• 

•a 
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\ 

>, 

/ 

25  — 

(  f.\ 

N 

SX^ 

c 

Unifon 

(A).    .. 
n  wmdinj 

^"N.^ 

"--^ 

^^ 

1? 

?•=> 

"^•^^ 

6  25 

0 
( 

8 

)            20            40            60           80           1C 

Per  cent  rotation  of  rheostat  shaft 

FIG.  21.    Typical  Curves  of  Current,  Voltage,  or 
Wattage  Relative  to  Shaft  Rotation  for  Uni- 
formly   Wound    and    Taper-wound    Rheostats 
(Courtesy  Ohmite  Mfg.  Co.) 


STEP-TYPE  POTENTIOMETERS,  RESISTANCE  BOXES,  AND  ATTENUATORS. 
Potentiometers.  This  type  consists  of  a  rotary-type  switch  wired  with  fixed  wire-wound 
resistors  between  successive  contact  positions.  A  brush  or  blade  rotated  by  a  shaft  makes 
contact  with  contact  studs  or  clips  to  which  the  fixed  resistances  are  wired.  The  resistors 
of  each  step  may  be  all  of  equal  resistance  value  or  they  may  differ,  depending  upon  the 
nature  of  the  application.  The  use  of  finite  resistance  steps  permits  a  high  degree  of  accu- 
racy, especially  in  low-  and  medium-frequency  applications.  The  accuracy  at  higher 
frequencies  is  dependent  upon  the  frequency  characteristics  of  the  individual  resistors  and 
the  type  of  switch  structure.  Likewise,  the  amount  of  power  that  can  be  dissipated  is 
governed  by  the  fixed  resistors  and  the  type  of  switch  used. 

Decade  Resistance  Boxes.  Many  laboratory  measurements  and  test  instruments  re- 
quire the  adjustment  of  resistance  in  a  circuit  in  accurately  known  steps  of  pure  resistance. 
A  decade  resistance  box  consists  of  a  number  of  individual  resistance  units  equipped  with 
switching  so  that  the  total  resistance  is  adjustable  in  decade  units.  Resistance  units  are 
connected  in  series,  and  as  many  units  can  be  connected  as  are  required.  To  keep  minimum 
inductance  and  distributed  capacitance,  card-wound  or  spool-type  resistor  units  are  used. 
Shielding  and  careful  wiring  arrangement  also  help  keep  capacitance  low  between  resistor 
units.  Where  residual  inductance  of  the  resistors  must  be  considered,  a  switching  arrange- 
ment introduces  a  compensating  winding  as  the  resistance  is  adjusted  to  maintain  constant 
inductance. 

Attenuators.  Attenuators  are  used  to  insert  known  amounts  of  transmission  loss  in 
circuits  either  for  testing  purposes  or  for  volume  level  control.  Step-type  attenuators 
basically  consist  of  rotary-type  switches  and  fixed  resistors,  as  in  step-type  potentiometers. 
They  are  arranged,  however,  to  introduce  various  types  of  balanced  or  unbalanced  resistive 
networks  into  a  circuit.  They  are  designed  electrically  to  be  inserted  between  specific 
input  and  output  impedances,  and  only  when  so  used  will  they  insert  the  desired  loss  (see 
Section  5). 

CARBON  COMPOSITION  TYPE  POTENTIOMETERS.  Carbon  composition  poten- 
tiometers are  widely  used  in  the  communications  and  electronics  fields  on  account  of  their 
low  cost,  the  higher  resistance  values  in  which  they  can  be  obtained,  and  their  excellent 
high-frequency  characteristics.  The  types  generally  available  are  physically  similar  to  the 
small  single-hole  mounting,  low-operating-temperature  wire-wound  controls  previously 
described.  Two  types  of  composition  resistance  elements  are  used,  namely,  the  film-coated 
type  and  the  molded  type.  In  the  film-coated  type,  the  carbon,  filler,  and  binder  mixture 


SPECIAL-PUBPOSE  RESISTORS 


3-21 


100 


are  applied  as  a  film  on  a  ring  of  insulating  material.  The  film  is  specially  processed  so  as 
to  minimize  abrasion  of  the  contact  surface  of  the  resistance  element.  In  the  molded  type 
the  carbon  composition  is  molded  into  a  phenolic  base.  The  contact  is  a  carbon  brush, 
giving  a  carbon  to  carbon  contact. 

Linear  and  non-linear  resistance  rotation  characteristics  are  obtainable  in  the  composi- 
tion types.  The  non-linear  or  tapered  characteristic  is  produced  by  varying  the  proportion 
of  the  conducting  material  to  the  insulating  material  in  the  mixture  as  the  element  pro- 
gresses over  its  length.  It  is  possible  by  blending  in  this  manner  to  obtain  a  rather  smooth 
rate  of  change  of  resistance  with  angular  rotation.  Typical  resistance-rotation  character- 
istics of  composition-type  potentiometers  are 
shown  in  Fig.  22.  Curve  A  represents  a  clock- 
wise linear  characteristic  except  for  a  small 
range  at  each  end  of  the  rotation.  Curve  B 
illustrates  a  clockwise  taper  in  which  the  first 
50  per  cent  of  the  rotation  introduces  only  10 
per  cent  of  the  resistance  into  the  circuit, 
Whereas  the  second  half  of  the  rotation  inserts 
the  remaining  90  per  cent  of  the  resistance. 
Curve  C  represents  a  clockwise  taper  in  which 
the  first  50  per  cent  of  the  rotation  introduces 
90  per  cent  of  the  resistance  and  the  second 
half  of  the  rotation  inserts  the  remaining  10 
per  cent  of  the  resistance.  Curves  D,  E,  and 
F  illustrate  counterclockwise  tapers  of  similar 
characteristics  to  curves  A,  B,  and  C,  respec- 
tively. 

Film-type  composition  potentiometers  are 
available  in  a  variety  of  sizes  ranging  from 
about  5/s  to  1  1/2  in.  in  diameter  and  in  watt- 
age ratings  from  about  0.05  watt  to  1  watt. 

The  available  molded  types  are  about  1  1/s  in.  in  diameter  and  are  rated  at  about  2  watts. 
Resistance  values  obtainable  range  from  about  50  ohms  to  10  megohms. 

For  film  and  molded  composition-type  potentiometers,  the  voltage  coefficient  and  the 
effects  of  overloading,  aging,  temperature  changes,  and  exposure  to  high  humidity  are 
about  of  the  same  order  as  for  film  and  molded  fixed  composition  resistors.  The  molded 
are  inherently  more  stable  in  resistance  than  the  film  types.  The  tabulation  below  ob- 
tained by  tests  compares  typical  film  and  molded  types  of  1-megohm  resistance  with 
respect  to  their  stability  of  resistance  under  varying  atmospheric  conditions.  The  tabu- 
lation is  in  terms  of  average  percentage  change  in  resistance  from  that  measured  at  room 
conditions  in  successive  tests  on  the  same  set  of  samples. 

Per  Cent  Change  in  Resistance  from  Initial  Resistance  at  20°  C  under  Varying  Atmos- 
pheric Conditions 


90 
|  70 

leo 
I  50 

'3  40 
c 
S  30 

$ 
^20 

10 
0 

^^N 

s 

V 

"^ 

^ 

f 

-~* 

"*"*" 

^ 

\ 

V 

\ 

/ 

\ 

/ 

^/ 

\ 

\ 

^  / 

\/ 

^ 

r 

-B 

E- 

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/s 

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y 

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\ 

/ 

v/ 

Cx 

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/ 

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-D 

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y 

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/**. 

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/ 

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-^ 

^ 

X- 

*^^ 

"\ 

\ 

0    10    20    30    40   50    60  70    80   90    100 
Per  cent  clockwise  shaft  rotation 

FIG.  22.     Nominal  Resistance-rotation  Char- 
acteristics (Courtesy  Ohmite  Mfg.  Co.) 


After  96  Hr 

After 

After 

After 

After 

at  40°  C 

After 

4  Hr  at 

4Hrat 

72  Hr  at 

72  Hr  at 

and  95% 

72hr  at 

-18°C 

-50°C 

50°  C 

65°  C 

relative 

90°  C 

humidity 

Film  type  

4-3.0 

+  7.0 

-7.0 

-10.0 

+  14.0 

-17.5 

Molded  type  

+  3.5 

+  5.5 

-4.0 

-5.5 

+  6.5 

-5.5 

Owing  to  their  low  inductance  and  capacitance,  composition  potentiometers  are  finding 
wide  use  in  high-frequency  applications;  an  example  is  a  common  shaft  tandem  arrange- 
ment of  three  rheostats,  two  of  which  are  the  series  arms  and  the  third  the  shunt  arm  of  a 
continuously  adjustable  T  type  attenuator  for  level  control  in  television  transmission  over 
telephone  circuits  at  frequencies  up  to  4  megacycles. 


7.  SPECIAL-PURPOSE  RESISTORS 

In  addition  to  the  many  resistors  described  so  far,  numerous  special-purpose  resistors 
designed  for  particular  applications  should  at  least  be  mentioned  here.  In  this  category 
are  resistor  standards  such  as  those  used  in  various  measuring  circuits  and  bridges  which 
are  discussed  in  Electrical  Measurements,  Section  11  of  this  handbook.  Thermistors  and 
varistors  are  used  in  an  increasing  number  of  applications  in  communications  circuits,  and 


3-22  RESISTORS,  INDUCTORS,    CAPACITORS 

they  are  fully  described  in  the  following  articles.  Resistors  used  primarily  for  heating  as 
in  furnace  elements  and  heavy-duty  power  controllers  are  covered  in  the  Electric  Power 
volume  of  the  Electrical  Engineers'  Handbook.  A  few  other  kinds  are: 

Resistance  Lamps.  In  this  type  a  special  lamp  filament  having  a  high  positive  tempera- 
ture coefficient  serves  either  as  a  current-limiting,  current-regulating,  or  protective  device 
in  communication  circuits.  Those  used  to  maintain  constant  current,  as  in  the  heater 
circuits  of  vacuum  tubes,  are  referred  to  as  ballast  lamps. 

Dummy  Antenna  Loads.  These  may  be  in  the  form  of  (a)  special  space-wound  resistance 
elements  enclosed  in  an  evacuated  glass  bulb,  (6)  a  wound  mica  card  mounted  between 
metal  castings  to  assist  in  carrying  off  the  heat,  or  (c)  a  water-cooled  film-type  resistor  in 
which  water  is  circulated  through  or  over  a  ceramic  core  on  which  a  resistor  film  has  been 
deposited. 

RESISTORS  IN  PRINTED  CIRCUITS.  Printed  circuit  structures  are  a  comparatively 
recent  development  in  which  a  network  of  resistors,  in  combination  with  capacitors  and 
inductors,  is  applied  in  the  form  of  bands  or  ribbon  to  one  or  both  sides  of  a  supporting 
structure,  usually  a  thin  ceramic  plate.  The  necessary  capacitors  are  also  mounted  on  the 
plate,  and  the  plate  itself  may  serve  as  the  dielectric  spacer  and  support  for  the  capacitor. 
Connections  to  both  resistors  and  capacitors  are  made  by  lines  of  conducting  paint.  Thus 
a  complete  coupling  or  filter  circuit  or  even  an  entire  amplifier  circuit  having  several 
stages  of  amplification  may  be  assembled  on  a  thin  plate  of  1  or  2  sq.  in.  of  surface  area. 
The  network  is  suitably  terminated  with  wire  leads,  and  the  whole  structure  is  given  a 
suitable  protective  covering.  The  resistors  may  be  of  metallic  or  carbon  film  or  of  the 
composition  film  type.  For  composition  and  metal  film  type  resistors  produced  by  chem- 
ical means,  the  material  may  be  applied  in  the  desired  pattern  by  painting  or  silk-screen 
process.  For  resistors  applied  by  the  metal-evaporation  process,  suitable  stencils  may  be 
used  to  limit  deposition  to  the  desired  areas.  Where  a  high  dielectric  body  serves  both  as 
the  dielectric  for  the  capacitors  in  the  circuit  and  as  the  supporting  panel  for  the  resistors, 
note  that  the  increased  distributed  capacitance  brought  about  in  the  resistor  due  to  the 
intimate  contact  with  the  high  dielectric  material  may  have  a  very  marked  effect  on  its 
performance  at  high  frequency.  Printed  circuits  are  finding  increasing  applications  in 
devices  such  as  hearing  aids,  miniature  radios,  or  wherever  space  is  at  a  premium.  Obvious 
advantages  are  compactness  and  small  size.  Disadvantages  are  difficulties  in  manufacture 
due  to  the  necessity  for  processing  all  elements  of  the  circuit  at  one  time,  which  makes 
close  tolerance  adjustment  of  individual  elements  impracticable  and  lessens  the  likelihood 
of  attaining  the  desired  characteristics  in  all  components.  Also,  in  service,  it  is  usually 
necessary  to  replace  the  entire  unit  when  a  single  element  becomes  defective. 

BIBLIOGRAPHY 

Industrial  Control  Standards,  Publication  1C4-22,  July,  1946,  National  Electrical  Manufacturers  Assoc., 

155  B.  44th  St.,  New  York,  N.  Y. 
Blackburn,  J.  P.,  Components  Handbook,  Vol.  17,  1949,  Radiation  Laboratory  Series,  Massachusetts 

Institute  of  Technology,  Cambridge,  Mass. 
Curtiss,  H.  L.,  and  Grover,  F.  W.  Resistance  Coils  for  A.C.  Work,  Bulletin  Bureau  of  Standards,  Vol.  8, 

pp.  495-517  (1913). 

B.  Hague,  Alternating  Current  Bridge  Methods,  3d  Edition,  Chapters  2,  3,  Pitman  &  Sons,  1932. 
Bureau  of  Standards  Circular  74. 

Bureau  of  Standards  Circular  100,  2d  Edition,  pp.  100-102. 
Wireless  Engineer,  Vol.  XII,  No.  141  (June,  1935);  No.  143  (August  1935). 
High-Frequency  Characteristics  of  Resistors,  Report  520,  Radiation  Laboratory,  Massachusetts  Institute 

of  Technology,  Cambridge,  Mass. 
Nettleton,  L.  A.,  and  Dole,  Fred  E.,  Potentiometers,  Review  of  Scientific  Instruments,  Vol.  17,  No.  10, 

pp.  356-363  (Oct.  19,  1946). 

Brunetti  and  Curtis,  Printed  Circuit  Techniques,  Bureau  of  Standards  Circular  468. 
New  Advances  in  Printed  Circuits,  Bureau  of  Standards,  Miscellaneous  Publication  192. 

Data  Sources: 

Driver-Harris  Co.,  Catalog  R-46. 
Wilbur  B.  Driver  Co.,  Resistance  Handbook. 
C.  O.  Jelliff  Mfr?.  Co.,  Resistance  Alloys. 
Hoskins  Mfg.  Co.,  Catalog  C. 


VARISTORS  AND  THERMISTORS 

By  N.  Y.  Priessman 

GENERAL.  The  term  varistor,  from  the  words  variable  and  resistor,  is  applied  to  a 
group  of  circuit  elements  broadly  classified  as  non-ohmic  resistances.  The  application 
of  the  term  varistor  is  restricted  to  devices  in  which  the  property  of  variable  resistance 
is  provided  by  solid  semiconductor  materials.  The  semiconductors  that  have  proved  use- 


COPPER-CUPROUS  OXIDE  VARISTOR 


3-23 


ful  as  stable  circuit  elements  are  those  in  which  the  current  carriers  are  electrons  as  dis- 
tinguished from  those  in  which  ions  are  transferred  through  the  solid. 

These  non-ohmic  resistors  may  be  divided  into  two  broad  classes,  depending  upon 
whether  the  resistance  change  is  an  electric  field  effect  (varistors)  or  a  temperature  effect 
(thermistors).  The  field-effect  varistors  may  further  be  divided  into  rectifier  varistors 
and  symmetrical  varistors.  Rectifiers,  such  as  copper  oxide,  selenium,  silicon,  and  ger- 
manium, exhibit  quite  different  values  of  resistance  depending  on  the  polarity  of  the 
applied  voltage.  Symmetrical  varistors  such  as  Thyrite,  Metrosil,  Atmite,  and  silicon 
carbide  show  no  rectifying  properties.  Thermistors  (see  article  10)  change  resistance 
markedly  with  changes  of  temperature  but  do  not,  independently  of  temperature  change, 
possess  a  non-linear  resistance  characteristic. 


0.0025"-0.0040" 


FIG.  1.     Copper  Oxide  Varistor  Cell 


8.  COPPER-CUPROUS  OXIDE  VARISTOR 

The  copper-cuprous  oxide  varistor  consists  essentially  of  a  piece  of  sheet  copper  about 
0.050  in.  thick  in  the  form  of  a  disk,  washer,  or  plate  which  has  been  oxidized  so  as  to  form 
on  its  surface  a  layer  of  red  cuprous  oxide.  A  thin  layer  of  conducting  material  is  applied 
to  the  exposed  surface  of  this  oxide  to  provide  a  contact,  known  as  the  outer  contact.  The 
mother  copper  provides  the  other  electrode.  Figure  1  shows  an  enlarged  and  out-of-scale 
cross-section  of  such  a  varistor  cell. 

The  manufacturing  techniques,  although  differing  in  detail  with  various  manufacturers, 
depending  on  the  particular  qualities  in  the  product  they  are  interested  in,  have  in  common 
the  following:  the  desired  form  of  cell  is  blanked 
from  sheet  copper  (usually  the  grade  known 
as  Chile  copper),  chemically  cleaned,  oxi- 
dized in  air  atmosphere  at  a  temperature  in 
the  neighborhood  of  1000  deg  cent  for  some 
10  to  20  minutes  to  form  a  layer  of  cuprous 
oxide  0.003  to  0.004  in.  thick,  held  for  some 
minutes  at  a  temperature  of  about  550  deg 
cent,  and  then  quenched  in  water.  At  this  stage  the  cuprous  oxide  (red)  is  covered  with 
a  thin  layer  of  cupric  oxide  (black)  which  must  be  removed  by  chemical  action.  The 
means  of  providing  an  outer  contact  vary  with  the  manufacturer.  The  following  means 
are  in  common  use:  (a)  painted-on  contacts  of  "aqua-dag"  (colloidal  graphite),  (b)  elec- 
troplated contacts,  and  (c)  contacts  of  gold  or  silver  produced  by  the  well-known  tech- 
niques of  evaporation  in  vacuum. 

The  electrical  properties  of  the  varistor  may  be  altered  significantly  by  suitable  varia- 
tions in  the  fabricating  processes,  as  for  example  by  the  addition  of  metallic  impurities  to 
the  copper  (such  as  thallium) ;  by  changes  in  time,  temperature,  and  atmosphere  of  the 
heat  treatments;  or  by  changes  in  rate  of  cooling  and  quench  temperature,  etc. 

The  current-voltage  characteristics  of  a  varistor  cell  commonly  used  in  communication 
circuits  is  shown  in  Fig.  2  in  the  "Chile  copper"  curves.  The  "forward  characteristic" 
exhibited  by  the  cell  with  the  mother  copper  negative  and  the  outer  contact  positive  is 
the  flatter  branch  of  the  two  curves.  The  "reverse  characteristic"  which  obtains  when  the 

copper  is  positive  and  the  outer  contact 
negative  is  shown  by  the  steeper  branch. 
Considering  in  detail  the  forward  charac- 
teristic curve  the  varistor  behaves  very 
much  like  an  ohmic  resistor  at  voltages 
below  0.05.  As  the  voltage  increases 
above  this  the  current  increases  rapidly; 
that  is,  the  resistance  decreases.  The  seat 
of  this  potential  dependent  resistance  as 
well  as  of  the  rectifying  property  is  in  the 
interface  between  the  cuprous  oxide  and 
the  copper.  This  interface  is  variously 
called  blocking  layer,  junction  layer,  bar- 
rier layer,  etc.  As  the  voltage  increases 
above  0.5  volt  the  current  increases  less 


FIG.  2.     1/2"  Diameter  Copper  Oxide  Varistor  Cell 
Representative  Voltage-current  Characteristics 


rapidly  and  the  resistance  approaches  a  limiting  constant  series  resistance,  which  is 
mainly  the  body  resistance  of  the  oxide  layer  itself.  The  series  resistance  of  the  outer 
contact  is  small  compared  with  the  body  resistance  of  the  cuprous  oxide  layer  and  is 
non-rectifying.  The  thallium-copper  cell  behaves  similarly  but  with  reduced  currents 
in  both  directions. 


3-24 


RESISTORS,  INDUCTORS,   CAPACITORS 


35 

~30 
S 
I  25 


§15 

$*~ 


\ 


0.01     0.050.1      0.5   1.          5  10         50100     5001000 
Time  of  application  in  seconds 

FIG.  3.    Delation  of  Safe  Reverse  Voltage  to  Time  of 

Application  for  Typical  Copper  Oxide  Cells  at  Ambient 

Temperatures    below    100°  F    and    for    Applications 

Spaced  at  Least  1  Minute  Apart 

practice.    Moreover  these  characteristics  change 
resistance  increases  with  time  and  temperature 


The  forward  current  at  a  voltage  is 
proportional  to  the  area  of  the  cell  or, 
more  accurately,  to  the  area  of  the 
outer  contact.  The  reverse  current  is 
not  simply  related  to  the  geometry  of 
the  cell.  The  characteristics  of  cells 
in  series  may  be  obtained  by  sliding 
the  characteristic  curves  parallel  to 
themselves  along  the  voltage  axis,  and 
the  characteristics  of  cells  in  parallel 
by  sliding  the  curves  along  the  current 
axis.  In  practice  large  areas  may  be 
obtained  most  economically  by  large- 
area  plates. 

Large  deviations  from  the  typical 
characteristic  curves  shown  are  inherent 
in  the  commercially  manufactured  prod- 
uct even  when  produced  under  the 
best  controlled  conditions  of  present 
with  time  and  use.  While  the  forward 
t,  the  reverse  resistance  decreases  with 


101 


10" 


icr 


10 


i 


10 


E       , 
o  10' 


10"9        10~8         1Q-7        1Q-*        10~5         10~4         10~3         10~2         10"1 
Current  in  amperes 

FIG.  4.     Representative  D-c  Characteristic  of  3/16-in.-diam.  Copper  Oxide  Varistor  Cell  (Chile  Copper) 

continued  application  of  reverse  voltage.    In  general  this  should  not  exceed  4  volts  (see 
Fig.  3) .    These  effects  again  vary  from  batch  to  batch. 

Large  negative  temperature  coefficients 
of  resistance  both  in  the  forward  and 
reverse  direction  are  characteristic  of  these 
devices.  No  simple  law  such  as  holds  for 
metallic  conductors  is  applicable,  and 
the  variation  of  resistance  with  tempera- 
ture may  best  be  displayed  graphically  as  in 
Fig.  4. 

The  effect  of  exposure  to  moisture  is  to 
reduce  the  reverse  resistance  of  all  types 
of  cells.  This  is  especially  troublesome  in 
the  small-diameter,  very-high-reverse-re- 
sistance cells.  The  forward  resistance  of 
cells  with  aqua-dag  contacts  is  increased 
by  exposure  to  moisture.  It  is  customary 
in  the  use  of  varistors  as  circuit  elements 
to  provide  substantial  moisture-proofing 
in  the  form  of  organic  coatings,  potting  in 
wax,  etc. 

Structures.  The  3/4-in.-diameter  and 
larger  cells  are  usually  made  in  the  form 
of  a  washer  and  clamped  together  with 
wiring  terminals  on  an  insulated  bolt. 
The  1/2-in.-diameter  and  smaller  cells  are 


+Volts 


FIG.  5.     Voltage  Limiter,  Two  l/2-in.-diam.  Cop- 
per Oxide  Varistor  Cells  (Copper  plus  Thallium) 


BIBLIOGRAPHY 


3-25 


FIG.  6.    Bridge  Connected  Modulator 


usually  assembled  with  wiring  terminals  and  pressure  spring  in  cylindrical  cavities  in 
insulating  blocks. 

Ratings.  When  copper  oxide  varistors  are  used  as  current-supply  rectifiers  their  ratings 
are  based  on  four  factors  on  which  definite  specifications  should  be  made,  with  allowance 
for  aging.  These  factors  are  (1)  the  d-c  voltage  and  current  output,  (2)  the  a-c  voltage 
input,  (3)  the  ambient  temperature,  and  (4)  the  means  for  cooling  the  varistor  cells. 

APPLICATIONS.  Voltage  Limiter.  It  may  be  seen  from  Fig.  2  that  as  the  voltage  in 
the  forward  direction  increases  from  0.1  to  1.0  volt  the  current  increases  more  than  1000 
fold.  A  varistor  may  be  connected  across 
the  input  terminals  of  a  network  to  act  as 
a  bypass  when  applied  voltages  are  sub- 
stantially above  the  normal  level.  Figure 
5  shows  the  resistance- voltage  characteristic 
of  such  a  varistor  in  which  two  V2-in.-diam- 
eter  cells  of  thallium  copper  (see  Fig.  2)  are 
connected  in  parallel  opposing  so  that  the 
combination  has  a  symmetrical  resistance- 
voltage  curve.  The  resistance  is  greater 
than  10,000  ohms  for  voltages  up  to  0.1  volt 
and  drops  to  about  5.0  ohms  at  1.0  volt.  As  the  voltage  increases  above  1.0  volt  the  resist- 
ance decreases  but  little;  it  is  being  limited  by  the  body  resistance  of  the  copper  oxide  layer. 
Modulator.  Copper  oxide  varistors  are  extensively  used  both  as  modulators  and  de- 
modulators in  carrier  telephone  systems.  A  bridge-connected  modulator  is  shown  in 
Fig.  6.  The  carrier  voltage  ec  is  made  large  compared  with  the  signal  voltage  ev.  When 
the  carrier  voltage  is  of  such  polarity  as  to  bias  the  varistors  in  the  forward  direction  they 
will  all  be  low  in  resistance,  offering  substantially  a  short  circuit  to  the  signal  current. 
When  the  carrier  voltage  reverses  the  varistors  all  have  a  high  reverse  resistance  and  the 
signal  current  appears  in  R%.  It  is  desirable  to  prevent  unmodulated  carrier  current  from 
appearing  in  R%,  and  this  is  done  by  selecting  the  four  varistors  in  the  arms  of  the  bridge 

to  have  voltage-current  characteristics  very  closely  alike 
so  that  the  bridge  is  balanced  throughout  the  cycle  of  carrier 
voltage.  (A  complete  discussion  of  the  varistor  modulator 
is  given  by  Caruthers,  Bell  System  Technical  Journal,  Vol. 
18,  315  [1939].) 

Control  of  Telephone  Relays.  Various  configurations  of 
relay  windings  and  varistors  may  be  used  to  modify  the 
response  characteristics  of  relays.  Several  such  arrange- 
ments are  shown  in  Fig.  7:  (a)  delayed  release  with  prac- 
tically no  effect  on  operate  time;  (6)  delayed  operate  with 
practically  no  effect  on  release  time.  Steady  operation  of 
relays  from  an  a-c  source  is  obviously  possible  using  the 
varistor  as  a  rectifier.  In  the  simple  arrangement  shown  in 
Fig.  7(c),  the  reverse  half-cycle  is  bypassed  through  the 
varistor  which  also  affords  a  path  for  the  slow  decay  of 
current  established  in  the  winding  during  the  previous  posi- 
tive half-cycle. 

Varistors  are  used  in  a  circuit  network  known  as  a 
"compandor" — a  contraction  of  the  words  "compressor" 
and  "expandor."  A  compressor  is  a  non-linear  transmission 
network  in  which  the  range  of  signal  power  output  is  com- 
pressed relative  to  the  range  of  power  input.  An  expandor  is  a  network  which  provides 
the  inverse  action.  Detailed  discussion  of  such  circuits,  also  called  "vario-lossers,"  may 
be  found  in  Bennett  and  Doba,  Trans.  A.I.E.E.,  Vol.  60,  17  (1941). 


(a) 


BIBLIOGRAPHY 

A  Bibliography  on  Metallic  Rectifiers  and  Their  Principal  Applications,  published  by  the  American 
Institute  of  Electrical  Engineering,  New  York,  N.  Y.  A  bibliography  of  more  than  500  items  on 
fundamental  theory,  various  types  of  rectifiers  and  their  applications. 

Bibliography  of  Literature  on  Rectifiers  and  Semi-conductors,  Royal  Aircraft  Establishment,  Issue  4, 
August  1946. 

H.  C.  Torrey  and  C.  A.  Whitman,  Crystal  Rectifiers,  Rad.  Lab.  Series,  McGraw-Hill  Book  Co, 


3-26 


RESISTORS,  INDUCTORS,   CAPACITORS 


9.  SILICON  CARBIDE  VARISTORS 


It  has  long  been  known  that  silicon  carbide  will,  under  suitable  conditions  of  contact, 
exhibit  a  non-linear  relationship  between  current  and  voltage.  This  may  readily  be 
demonstrated  by  measuring  the  voltage-current  characteristic  of  a  mass  of  small  particles 
of  silicon  carbide  compressed  between  metallic  electrodes.  As  the  voltage  is  increased 
from  zero  the  current  increases,  at  low  voltage  in  direct  proportion  to  the  voltage  and  then 
much  more  rapidly.  If  the  number  of  particles  in  the  mass  is  large  and  the  distance  be- 
tween electrodes  large  compared  with  the  dimensions  of  the  particles  the  non-linear  resist- 
ance of  the  device  is  independent  of  polarity. 

Experiments  upon  single  particles  with  suitably  made  contacts  indicate  that  the  body 
resistance  of  the  particle  is  small,  ohmic,  and  independent  of  polarity. 

The  non-linear  conduction  exhibited  by  the  mass  of  particles  results  from  the  voltage- 
dependent  resistances  at  the  point-to-point  contacts  between  the  granules  of  silicon  carbide. 
The  overall  resistance  characteristic  may  be  thought  of  as  due  to  large  numbers  of  non- 
linear resistance  contacts  arranged  at  random  in  series  and  parallel.  In  a  statistical  sense 
the  aggregate  displays  no  dependence  upon  the  direction  of  current  flow.  This  varistor 
is  an  example  of  a  "symmetrical  non-linear  resistor." 

The  simple  device  of  containing  a  mass  of  silicon  carbide  particles  under  pressure  be- 
tween electrodes  does  not  have  the  stability  of  characteristic  under  use  conditions  to  afford 
wholly  reliable  circuit  elements. 

In  1930,  McEachron  (see  Journal  A.I.E.E.,  Vol.  49,  410  [1930])  described  a  silicon 
carbide  ceramic  non-linear  resistor  to  which  the  name  Thyrite  was  given.  The  material 
consists  of  silicon  carbide  particles  bonded  in  a  ceramic  matrix.  Similar  materials  are 
known  under  various  names  such  as  Metrosil  and  Atmite. 

The  essential  steps  of  manufacture  are  these:  suitable  silicon  carbide  particles,  clay  and 
water,  sometimes  with  a  minor  constituent  such  as  carbon,  are  mixed  to  form  a  plastic 
mass.  The  mass  is  partially  dried  and  forced  through  screens  to  obtain  a  slightly  damp 
granular  powder.  This  material  is  compressed  under  high  pressure  into  desired  shapes, 
generally  flat  disks  or  rods.  These  pieces  are  further  dried  and  heat  treated  in  a  reducing 
atmosphere  at  a  temperature  in  the  neighborhood  of  1200  deg  cent.  The  fired  pieces  are 
hard  and  strong  and  have  mechanical  properties  quite  similar  to  those  of  dry  process 
porcelain.  Electrodes  on  the  opposite  plane  faces  are  provided  by  spraying  or  Schooping 
a  layer  of  metal  such  as  brass,  copper,  aluminum,  or  tin.  The  piece  is  then  usually  im- 
pregnated with  a  moisture-repellent  organic  substance  to  prevent  pickup  of  water,  which 
adversely  affects  their  electrical  stability. 

The  electrical  properties  of  the  product  are  profoundly  affected  by  the  parameters  of 
process:  materials,  particle  size,  moisture  content,  forming  pressure,  and  especially  tem- 
perature, time,  and  atmosphere  of  the  heat  treatments.  The  products  of  different  manu- 

facturers  differ  somewhat  in  electrical 
properties,  most  importantly  in  the  de- 
gree of  non-linearity,  and  the  character- 
istics of  Fig.  8  are  to  be  taken  only  as 
generally  indicative.  The  current-volt- 
age characteristic  shown  is  closely  rep- 
resented by  the  equation 


1000, 


Co'ntinuou's 
operation  ' 


intermittent 
operation 


10'3  ICT 

Amperes 


10" 


FIG.  8.     Representative  D-c  Characteristics  of  Some 
3/4-in.-diam.  Silicon  Carbide  Varistor  Disks 


where  I  =  current  through  the  piece, 
E  =  voltage  applied  to  the  piece,  Ci  and 
Cz  are  constants  depending  on  the  mate- 
rial and  geometry  of  the  piece,  and  n  is 
an  exponent  the  value  of  which  depends 
on  various  factors  in  the  manufacturing 
process  and  generally  lies  between  3.5 
and  5.0.  Some  manufacturers  indicate 
values  of  n  as  high  as  7.0  but  only  for 
pieces  having  resistances  much  above 
the  range  indicated  in  Fig.  8. 


The  variation  of  characteristic  through  control  of  manufacturing  processes  and  geometry 
of  the  piece  permits  coverage  of  an  enormous  range  of  current  and  voltage.  This  range 
may  be  further  extended  by  connection  of  pieces  in  series  or  parallel.  It  is  to  be  noted 
from  Fig.  8  that  as  the  resistance  of  the  piece  decreases  the  value  of  n  decreases  also, 'and 
this  being  typical  of  aU  manufacturers7  products  may  be  considered  an  inherent  charac- 


SILICON  CARBIDE  VARISTOKS 


3-27 


teristic  of  the  presently  made  material.  In  consequence  it  is  not  possible  with  this  device 
to  obtain  marked  non-linearity  at  low  voltage. 

In  common  with  semiconductors  the  silicon  carbide  varistor  exhibits  a  negative  tem- 
perature coefficient  of  resistance.  The  coefficient  does  not  have  a  single  value  but  varies 
both  with  the  material  and  with  voltage  and  temperature.  The  values  of  the  coefficient 
at  constant  voltage  cover  a  spread  of  from  0.3  per  cent  to  0.9  per  cent  per  degree  centigrade 
in  the  normally  used  range  of  temperature.  The  higher  values  of  temperature  coefficient 
are  observed  at  the  lower  voltages. 

At  high  frequencies  consideration  should  be  given  to  the  presence  of  a  capacitance 
effectively  in  parallel  with  the  non-ohmic  resistance.  The  exact  value  of  this  capacitance 
is  determinable  only  by  measurement,  but  the  order  of  magnitude  may  be  calculated  by 
assuming  the  material  to  have  a  dielectric  constant  of  30  to  200. 

Commonly  used  shapes  are  rods  and  disks.  Small  disks  and  rods  may  be  furnished  with 
leads  soldered  to  the  metallic  electrodes  on  the  faces  of  the  piece.  Disks  are  also  made 
with  holes  in  the  center  and  clamped  together  with  wiring  terminals  by  means  of  a  central 
bolt.  Disks  and  rods  of  all  sizes  are  used  with  spring  clip  mountings  which  furnish  mechan- 
ical support  and  electrical  connections. 

"When  used  under  high  humidity  conditions,  or  at  low  currents,  the  organic  impregnant, 
referred  to  in  the  description  of  the  fabricating  process,  may  not  be  sufficient  protection 
against  moisture  and  further  precautions  may  be  necessary. 

Approximate  values  of  mechanical  and  thermal  properties  of  importance  in  circuit 
element  design  are  as  follows: 

Bulk  density 2.35  grams  per  cu  cm 

Compression  strength 15,000  to  23,000  Ib  per  sq  in 

Specific  heat 0.17  to  0.21  cal  per  gram  per  deg  cent 

Thermal  conductivity 0.0034  cal  per  cm  per  sec  per  deg  cent 

Requirements  on  the  current-voltage  characteristic  for  a  particular  application  may  be 
stated  in  a  number  of  ways;  the  following  are  commonly  used. 

(a)  The  voltage  J$\  at  a  current  Ji  shall  be  greater  than  some  value,  and  the  voltage  E% 
at  a  current  /a,  where  /2  is  greater  than  Ji,  shall  be  less  than  some  value.  This  statement 
of  requirements  contains  implicitly  a  requirement  as  to  the  minimum  value  of  n. 

(6)  The  voltage  at  a  given  current  I  shall  be  equal  to  a  value  E  ±  X  per  cent,  and  the 
value  of  n  shall  lie  within  certain  limits  throughout  a  range  of  current. 

It  is  to  be  noted  that  considerable  differences  in  characteristic  may  exist  between  pieces 
meeting  a  set  of  such  requirements.  In  commercial  manufacture  the  range  of  voltage  at  a 
given  current  commonly  runs  ±20%  about  the  average.  Accuracy  of  meters  used  in 
checking  requirements  is  im- 
portant since  errors  in  volt- 
age readings  are  to  be  multi- 
plied by  n  in  determining 
their  effect  on  current  read- 
ings. 

Self-heating  resulting  from 
power  dissipation  in  the  var- 
istor lowers  its  resistance 
(negative  temperature  coef- 
ficient of  resistance) ,  but  this 
effect  is  in  general  reversible; 
that  is,  no  permanent  effects 
on  the  characteristic  are  pro- 
duced by  moderate  heating, 
say  from  100  to  150  deg  cent. 
The  safe  upper  limit  of  heat- 
ing is  oftentimes  determined 
by  the  moisture-resistant  or- 
ganic compound  used  as  an 
impregnant.  As  shown  in 
Fig.  8,  1.0  watt  for  a  disk  of 
3/4-in.  diameter  suspended  in 
free  air  at  50  deg  cent  is  a  limit  recommended  by  one  manufacturer.  Very  heavy  transient 
currents  may  alter  permanently  the  characteristic,  usually  in  the  direction  of  decreasing 
the  resistance. 

APPLICATIONS.  (1)  A  silicon  carbide  varistor  connected  across  the  terminals  of  an 
electromagnetic  winding  acts  to  limit  the  surge  voltage  generated  when  the  field  is  opened, 


Surge  voltage  prote 
by  use  of  silicon  ca 
vanster 


0.05   0.1 


5      10 


30  50     100 


0.5      1, 

Milliamperes 
FIG.  9.    Surge  Voltage  Protection  by  Use  of  Silicon  Carbide  Varistor 


3-28  RESISTORS,   INDUCTORS^   CAPACITORS 

As  shown  in  Fig.  9  the  maximum  value  of  voltage  across  the  varistor  may  be  determined 
from  the  point  on  the  voltage-current  characteristic  corresponding  to  the  steady-state 
value  of  current  IQ  in  the  winding.  As  compared  with  an  ordinary  resistance  shunt  across 
the  winding  to  secure  the  same  voltage-limiting  effect,  the  varistor  dissipates  much  less 
power  when  the  coil  is  steadily  energized, 

(2)  In  certain  carrier  telephone  system  filters  exposed  to  high  incoming  voltage,  the 
condenser  of  a  bigh-Q  combination  of  coil  and  condenser  has  been  protected  by  a  varistor 
in  shunt. 

(3)  Some  of  the  smaller  telephone  switchboards  have  line  lamps  connected  directly  in 
the  subscriber's  loop  for  signaling.    These  line  lamps  are  exposed  to  electrical  disturbances 
that  may  be  impressed  on  the  outside  lines,  and  if  the  disturbances  are  severe  enough  the 
lamps  may  be  burned  out.    Silicon  carbide  varistors  have  been  used  very  effectively  in 
parallel  with  the  lamp  to  bypass  large  incoming  surges.    The  high  resistance  of  the  varistor 
at  the  normal  signaling  level  has  no  appreciable  effect  on  the  lamp  illumination. 

(4)  Use  is  made  of  varistors  to  protect  contacts  controlling  inductive  circuits  from  the 
deleterious  effect  of  sparks  resulting  from  the  opening  of  such  circuits.    Usually  the  varistor 
is  connected  across  the  winding  rather  than  across  the  contact  to  avoid  continuous  current 
drain.    Though  such  an  arrangement  is  useful  it  is  not  a  satisfactory  general  solution  of  the 
problem.    The  varistor  increases  the  release  time  of  the  relay  or  switch  magnet,  though 
not  to  the  extent  that  an  ohmic  resistance  of  equivalent  spark  quenching  action  would  do, 
and  it  does  not  entirely  eliminate  high-frequency  oscillations  across  the  opening  contact 
due  to  the  associated  wiring. 

BIBLIOGRAPHY 

McEachron,  J.  A.I.E,E.,  Vol.  49,  410  (1930).     (Thyrite.) 

Fairweather,  J.  A.I.E.E.,  Pt.  I,  Vol.  89,  499  (1942).     (General.    Contains  bibliography  of  59  items.) 

Ash  worth,  Needham,  and  Sillars,  J.  A.I.E.E.,  Pt.  I,  Vol.  93,  385  (1946).  (General.  Contains  bibli- 
ography of  25  items.) 

Brownlee,  Gen.  Elec.  Rev..  Vol.  37,  175  (1934);  Vol.  37,  218  (1934).     (Calculation  of  circuits.) 

Grisdale,  Bell  Labs.  Record,  Vol.  19,  153  (1941).    (General.) 

Vigren,  Telegrafstyrelsen,  No.  7,  Vol.  9,  138  (1942).     (Contact  protection.) 

Royal  Aircraft  Establishment  Issue  4,  August  1946,  Bibliography  of  Literature  on  Rectifiers  and  Semi- 
conductors. 

10.  THERMISTORS 

Thermistors  or  thermally  sensitive  resistances  are  devices  made  of  solid  semiconductors 
the  electrical  resistance  of  which  varies  markedly  with  temperature.  This  phenomenon 
has  long  been  known,  Faraday  having  observed  that  the  resistance  of  silver  sulfide  de- 
creased rapidly  as  the  temperature  increased.  Since  that  time  it  has  been  determined  that 
a  great  number  of  materials  classed  electrically  as  semiconductors  exhibit  high  negative 
temperature  coefficients  of  resistance.  Semiconductors  have  specific  resistances  at  room 
temperature  much  greater  than  those  of  metallic  conductors  and  much  less  than  those  of 
insulators.  This  very  wide  intermediate  range  of  resistivities  is  not  bounded  precisely 
but  may  extend  from  0.1  ohm  cm  to  109  ohm  cm.  Materials  commercially  employed  in 
thermistor  circuit  elements  have  a  much  narrower  range  of  resistivity,  roughly  from 
10  ohm  cms  to  100,000  ohm  cm. 

The  materials  of  thermistor  construction  include  a  wide  variety  of  metallic  oxides.  In 
common  use  are  the  oxides  of  uranium  and  various  mixtures  of  the  oxides  of  magnesium, 
manganese,  titanium,  iron,  nickel,  cobalt,  zinc,  etc.  The  common  method  of  fabricating 
is  to  heat  the  oxides  in  the  form  of  compressed  powders  to  a  temperature  at  which  they  will 
sinter.  At  the  sintering  temperature  the  powders  recrystallize  to  form  a  dense,  hard, 
ceramic-like  solid  of  homogeneous  composition.  The  sintered-powder  process  permits  the 
mixing  of  various  oxides  in  suitable  proportions  to  produce  a  wide  range  of  electrical  and 
thermal  characteristics  and  permits  as  well  the  fabrication  of  a  great  variety  of  shapes  and 
sizes  of  the  completed  piece. 

Forms.  Three  forms  of  thermistors  are  common — disks,  rods,  and  beads.  A  thin  plate 
or  flake  form  has  also  been  described  and  is  hi  limited  use  (see  Becker  et  al.,  Bell  Sys.  Tech. 
Jour.,  January  1947).  Disks  range  in  diameter  from  0.125  to  2.0  in.  and  in  thickness  from 
0.030  to  0.250  in.  Rods  are  made  in  diameter  from  0.030  to  0.250  in.  and  in  length  from 
0.050  to  2.5  in.  Bead  diameters  range  from  0.006  to  0.060  in. 

Properties.  The  relations  between  specific  resistance  and  temperature  of  several 
thermistor  materials  are  shown  in  Fig.  10  and  for  comparison  the  resistance-temperature 
relation  of  platinum.  In  Fig.  11  the  log  of  the  specific  resistance  is  plotted  against  the 
reciprocal  of  the  absolute  temperature.  It  is  seen  that  the  curves  are  very  nearly  straight 
lines,  and  so  to  a  close  approximation 


THERMISTOES 


3-29 


log  p 


const.  +  j8  •  —     or 


const. 


from  which. 


where  T  =  temperature  in  degrees  Kelvin,  p  —  po  when  T  —  To,  0  is  numerically  propor- 
tional to  the  slope  and  is  of  the  dimensions  degrees  Kelvin.  From  the  definition  of  tem- 
perature coefficient  a.  =  (l/R)(dR/dT}  it  is  7 
seen  that  c 


10s 

10' 

E 
o 

I  10* 

c 

8    10° 


Crt 


2  10  ' 
10" 
10" 


•S 

*io* 


Iff 


io~lLH 


100        0        100      200      300     400 
Temperature  in  degrees  centigrade 

FIG.  10 


1.0 


2.0 


3.0          4,0 

1 


Temperature  in  degrees  Kelvin 
FIG.  11 


THERMISTOR  APPLICATIONS.  Direct  utilization  of  the  resistance-temperature 
relation  is  a  broad  field  of  use  including  resistance  thermometry,  compensation  for  the 
positive  temperature  coefficient  of  other  resistive  circuit  elements,  temperature  control, 
and  the  like.  In  all  these  applications  the  self-heating  effect  of  any  current  in  the  thermis- 
tor is  kept  small  so  that  the  resistance  is  fully  controlled  by  the  ambient  temperature. 

The  equation  previously  given  for  the  relation  between  resistance  and  temperature  may 
for  purposes  of  general  calculation  be 

Table  9.     Temperature-resistance  Characteris- 
tic of  a  Typical  Thermistor  Thermometer 


considered  independent  of  temperature. 
Table  9  shows  temperature-resistance 
characteristics  of  a  typical  thermistor 
thermometer.  With  an  ordinary  Wheat- 
stone  bridge  and  galvanometer  and  a 
suitably  calibrated  thermistor  thermom- 
eter a  precision  of  0.001  cent  deg  is 
readily  obtainable. 

The  use  of  thermistors  in  conjunction 
with  relays,  valves,  etc.,  for  temperature 
control  is  closely  akin  to  their  use  in 
thermometry.  The  larger  currents  re- 
quired for  relay  operation  necessitate 
design  consideration  of  the  self-heating 
effects  in  the  thermistor. 

Thermistors  are  used  to  compensate 
for  changes  in  resistance  of  electrical 
circuits  caused  by  ambient  temperature 
variations.  Shunting  the  thermistor  by 
a  parallel  resistance  sometimes  improves 
the  accuracy  of  the  compensation.  Consideration  should  be  given  to  like  temperature 
exposure  of  the  thermistor  and  the  compensated  circuit  element,  and  also  to  the  effects 
on  both  of  power  dissipation. 

Small  thermistors  have  been  used  extensively  to  measure  power  in  very  high-frequency 
test  sets.  Suitably  mounted  in  a  properly  terminated  waveguide  a  thermistor  bead  absorbs 
effectively  the  entering  power,  and  the  consequent  heating  of  the  bead  produces  a  change 


Temperature, 
deg  cent 

Resistance, 
ohms 

Temperature  Coefficients 

B, 

deg  cent 

a,  per  cent 
per  deg  cent 

-25 
0 
25 
50 
75 
100 
150 
200 

580,000 
145,000 
46,000 
16,400 
6,700 
3,200 
830 
305 

3,780 
3,850 
3,920 
3,980 
4,050 
4,120 
4,260 
4,410 

-6.  1 
-5.2 
-4.4 
-3.8 
-3.3 
-3.0 
-2.4 
-2.0 

Dissipation  constant  in  still  air,  approximately  4 
milliwatts  per  degree  centigrade;  thermal  time  constant 
in  still  air,  approximately  70  sec;  dimensions  of  thermis- 
tor, diameter  approximately  0.11  in.,  length  approxi- 
mately 0.54  in. 


3-30 


RESISTORS^   INDUCTORS,    CAPACITORS 


100 


in  resistance  which  may  readily  be  measured  with  high  accuracy.     Calibrating  may  be 
done  with  d-c  or  low-frequency  power. 

The  self-heating  effect  of  current  through  a  thermistor,  primarily  the  bead  type,  results 

in  interesting  and  useful  non-linear 
relationships  between  current  and 
voltage.  Figure  12  shows  a  "steady- 
state"  characteristic  of  a  particular 
form  of  bead  thermistor.  At  small 
currents  the  power  dissipated  is  too 
small  to  heat  the  thermistor  appre- 
ciably,  and  the  resistance  remains 
constant.  With  increasing  current  the 
°a  effects  of  self-heating  become  evident; 

\  the  temperature  of  the  thermistor  rises 
and  the  resistance  decreases.  As  the 
current  continues  to  increase,  the  slope 
of  the  curve  changes  from  positive  to 
negative  and  in  this  latter  region  the 
thermistor  exhibits  a  negative  value  of 
dv/dl,  that  is,  a  negative  resistance. 
The  numbers  along  the  curve  give  the 
rise  in  temperature  in  degrees  centi- 
!  grade  above  the  ambient.  The  change 

in  resistance  of  the  thermistor  does  not  occur  instantaneously  with  current  change  because 
of  its  thermal  mass. 

The  heating  and  consequent  reduction  of  resistance  by  the  continued  passage  of  current 
is  used  to  obtain  delayed  response  circuits  as  well  as  non-response  to  short-duration  surges 

by  connection  of  the  thermistor  in  series  with  a 
relay. 

Figure  13  shows  a  combination  of  thermistor 
and  resistances  to  obtain  either  a  speech  volume 
limiter  or  a  volume  compressor.  The  speed  of 
response  of  the  thermistor  is  adjusted  to  syllabic 
frequency  or  slower  to  eliminate  the  wave-form 
distortion  and  peak  chopping  common  to  quick- 
acting  non-linear  devices. 


100 


.LLm  i  tejLO  c_co  m  p  r  ess  o 


Load 


Directly  heated 
"    thermistor 


FIG.  14 


4          8         12        16 
Current  in  miliiamperes 

FIG.  13 


20 


Curve   1,   thermistor  characteristic 
Curves  2  and  4,  ohmic  resistor  char- 
acteristics 

Curves  3  and  5,  combined  charac- 
teristics 


Figure  14  shows  a  thermistor  in  the  negative 
feedback  circuit  of  an  amplifier  to  obtain  con- 
stant level  output  independent  of  variations  of 
signal  input.  This  use  is  of  great  importance  in 
telephone  carrier  systems  to  correct  for  variations 
in  overall  line  loss.  The  simple  bead  structure 
previously  described  is  not  adequate  for  this  pur- 
pose since  its  resistance  and  hence  the  amount  of 
feedback  would  be  subject  to  change  with  changes 
in  ambient  temperature.  Temperature  compen- 


sation is  economically  obtained  by  associating  a  heater  winding  with  the  bead  and  regulat- 
ing the  input  current  to  the  heater  to  produce  a  constant  temperature  surrounding  the  bead. 


BIBLIOGRAPHY 

1.  Pearson,  G.  L-,  Thermistors:  Their  Characteristics  and  Uses,  Bell.  Labs.  Rec.,  November  1940, 
.  85. 


.        .,  , 

p.  85. 

2.  SiUars,  R.  W     Materials  and  Devices  of  Falling  Resistance-Temperature  Characteristics,  J.  Sci. 
InsL,  Vol.  19,  No.  6,  81  (June  1942). 

££'  T-  r8^  ^  V*n  Iima'  Ml'  Thermistors,  Their  Characteristics,  Uses  and  Associated  Circuits, 
Office  of  Publication  Board  Report  PB-3407,  1945,  Dept.  of  Commerce,  Washington  25  D   C 


PKOPERTIES  OF  AIR-CORE  INDUCTORS  3-31 

4.  Gray,  T.  S.,  and  Van  Dilla,  M.,  A  Thermistor  Electronic  Thermoregulator,  Office  of  Publication 

Board  Report  BP-3402  1945,  Dept.  of  Commerce,  Washington  25,  D.  C. 

5.  Gaffney,  F.  J.,  Microwave  Measurements  and  Test  Equipments,  Proc.  I.R.E.,  Vol.  34,  775-793 

(October  1946). 

6.  Becker,  J.  A.,  Green,  C.  BM  and  Pearson,  G.  L.,  Properties  and  Uses  of  Thermistors — Thermally 

Sensitive  Resistors,  Electrical  Engineering,  Vol.  65,  November  1946,  and  Bell  Sys.  Tech.  J.,  Vol. 
26,  No.  1,  170-212  (January  1947). 

7.  Pearson,  G.  L.,  The  Physics  of  Electronic  Semi-conductors,  Trans.  A.I.E.E.,  Vol.  66,  209  (1947). 

8.  Roloff,  C.  C.,  Thermo-variable  Resistors,  Elec.  Rev.  Lpnd.,  Vol.  140,  315  (February  1947). 

9.  Verwey,  Haayman,  and  Romeyn,  Semi-conductors  with  Large  Negative  Temperature  Coefficient 

of  Resistance,  Phillips  Tech.  Rev.,  Vol.  9,  No.  8,  239  (1947). 

10.  Montgomery,  C.  G.,  Technique  of  Microwave  Measurements,  M.I.T.  Rad.  Lab.  Series.     (Use  of 

thermistors,  Chapter  3.) 

11.  Deeter,  E.  L.,  Null  Temperature  Bridge,  Electronics,  Vol.  21,  No.  5,  180  (May  1948). 


INDUCTORS  WITH  AIR  CORES 

By  L.  M.  Hershey 

Of  the  many  different  types  of  air-core  inductors  in  communications  equipment  today, 
the  solenoid  and  the  universal  winding  are  most  widely  used.  Torroidal  windings,  and 
other  types  such  as  the  bank  winding,  spiral  winding,  and  basket  weave,  are  sometimes 
found. 

The  single-layer  solenoid  is  used  in  untuned  or  tuned  circuits,  for  resonant  circuits, 
chokes,  and  in  various  other  applications  where  a  high  Q,  low  distributed  capacitance, 
mechanical  strength,  or  ease  of  construction  is  of  importance.  More  space  is  required  to 
accommodate  a  single-layer  solenoid  than  a  universal  winding  of  any  type,  for  a  given 
inductance;  at  frequencies  below  about  1  or  2  me,  space  considerations  frequently  prevent 
the  use  of  the  single-layer  solenoid. 

The  universal  winding  is  popular  in  applications  similar  to  those  listed  above  for  the 
single-layer  solenoid  but  generally  at  frequencies  below  about  2  me.  It  produces  a  coil 
having  fairly  high  Q  with  low  distributed  capacitance,  and  mechanical  strength.  The 
distributed  capacitance  of  a  universal  winding  can  be  decreased  by  winding  it  on  a  narrower 
cam  or  by  building  it  in  a  number  of  sections,  each  of  which  is  a  universal  winding,  with 
these  sections  connected  in  series.  This  multisection  universal  winding  also  can  be  ad- 
justed to  close  inductance  tolerance  requirements  by  moving  one  of  the  end  sections  (some- 
times called  a  "pi")  nearer  to,  or  away  from,  the  adjacent  section. 

Many  special  types  of  windings  are  in  use  as  tuned  loops  in  the  broadcast  band  (540  to 
1600  kc).  Among  these  types  are  the  multilayer  solenoid,  basket  weave,  single-layer 
solenoid,  and  spiral  winding. 

Air-core  inductors  are  used  at  frequencies  up  to  about  200  me,  or  up  to  a  frequency 
where  a  transmission  line  becomes  more  convenient.  The  transition  region  between  coils 
and  transmission  lines  appears  to  be  extremely  broad. 

The  choice  of  the  type  of  inductor  is  generally  dictated  by  such  practical  considerations 
as  available  space  or  cost  as  well  as  by  circuit  considerations.  Untuned  primary  coils 
and  chokes  of  less  than  about  10  to  20  ^h  are  usually  solenoid  windings.  When  greater 
inductance  is  required,  the  universal  winding  is  used. 

11.  PROPERTIES  OF  AIR-CORE  INDUCTORS 

FIGURE  OF  MERIT.  The  figure  of  merit  of  an  inductor  is  the  ratio  of  its  effective 
reactance  to  its  effective  resistance.  This  factor  is  called  the  Q  of  the  inductor. 

POWER  FACTOR.  The  reciprocal  of  the  Q  of  the  inductor  is  equal  to  the  power  factor 
of  the  circuit  within  very  close  limits  for  values  of  Q  above  about  20.  The  power  factor 
is  more  convenient  than  the  Q  to  use  in  the  calculation  of  certain  circuit  phenomena.  For 
instance,  the  power  factor  of  a  circuit  formed  by  an  inductor  shunted  by  a  capacitor  is 
the  sum  of  the  power  factors  of  the  two  branches  of  the  circuit,  while  the  Q  of  the  circuit 
is  the  reciprocal  of  the  sum  of  the  reciprocals  of  the  Q's  of  each  branch. 

Like  resistance  and  conductance,  both  the  power  factor  and  the  Q  are  useful  concepts, 
and  the  choice  depends  upon  their  application  to  the  particular  problem. 

TIME  CONSTANT.  The  time  constant  of  an  inductor  in  series  with  a  resistor  (the 
resistor  may  represent  the  internal  resistance  of  the  inductor)  is  L/R;  it  is  the  time  in 
seconds  required  for  the  current,  through  an  inductor  of  L  henries  in  series  with  a  resistor 
of  R  ohms,  to  reach  0.632  of  its  final  value  if  a  voltage  is  applied  suddenly,  or  for  the  current 
through  the  series  circuit  to  fall  to  0.368  of  its  initial  value  if  the  inductor  and  resistor  are 
short-circuited  suddenly. 


3-32  RESISTORS,  INDUCTORS,   CAPACITORS 

COIL  LOSSES.  The  principal  losses  in  an  air-core  inductor  are  those  due  to  PR  loss 
in  the  conductor  and  the  dielectric  losses  in  the  coil  form,  wire  insulation,  impregnating 
material,  etc.  Eddy-current  losses  also  occur  in  the  conductor.  At  very  high  frequencies 
the  losses  due  to  radiated  power  may  be  appreciable.  Additional  losses  occur  outside  the 
coil  itself  whenever  any  magnetic,  dielectric,  or  conducting  material  is  within  the  field  of 
the  coil. 

The  attainment  of  a  maximum  Q  in  a  given  space  is  one  of  the  most  common  problems. 
At  the  lowest  frequencies,  the  problem  is  to  obtain  the  lowest  d-c  resistance  for  a  given 
inductance.  As  the  frequency  becomes  higher,  skin  effect  (the  tendency  of  an  alternating 
current  to  flow  along  the  outside  surface  of  a  conductor)  becomes  apparent,  and  is  measur- 
able even  at  power-line  frequencies.  Dielectric  losses  also  begin  to  be  noticeable  at  very 
low  frequencies.  Throughout  the  radio-frequency  spectrum  these  two  causes  of  power  loss 
are  of  extreme  importance.  For  example,  a  well-designed  coil  operating  at  1  me  might 
have  a  d-c  resistance  of  less  than  5  ohms  and  an  apparent  resistance  greater  than  10  ohms. 

Coil  losses  may  be  minimized,  and  high  values  of  Q  realized,  by  careful  choice  of  the  type 
of  wire  (to  increase  the  surface  area  of  the  conductor  through  which  the  r-f  currents  flow) , 
by  obtaining  the  optimum  spacing  between  conductors,  and  by  choosing  the  proper  shape 
of  the  winding  for  the  available  space. 

It  is  sometimes  desirable  to  design  a  coil  with  a  certain  value  of  Q  so  that,  for  instance,  a 
required  band  width  can  be  produced  without  the  use  of  an  external  damping  resistance. 
Then  the  designer  may  reverse  his  usual  thought  processes  and  use  a  "poor"  shape  factor 
for  his  coil,  a  conductor  either  larger  or  smaller  than  the  optimum  value  for  maximum  Q, 
or  use  a  value  other  than  optimum  for  the  spacing  between  conductors.  Frequently,  the 
diameter  of  the  coil  form  can  be  reduced  until  a  desired  low  value  of  Q  is  realized. 

DISTRIBUTED  CAPACITANCE.  Each  turn  of  an  inductor  is  coupled  magnetically 
to  other  turns  of  the  same  inductor.  A  certain  small  amount  of  capacitance  between  the 
turns  of  a  winding  is  unavoidably  produced  by  their  proximity.  The  effect  of  all  these 
small  series  capacitances  across  the  whole  inductor  at  its  working  frequency  is  called  the 
"distributed  capacitance  of  the  coil.  The  values  of  the  distributed  capacitances  of  various 
common  types  of  air-core  inductors  range  from  a  small  fraction  of  a  micro-microfarad  up 
to  10  or  more  ju/if.  In  general,  a  coil  having  a  large  ratio  of  length  to  diameter  has  a  low 
distributed  capacitance,  and  it  is  obvious  that  finer  wire  and  greater  spacing  between  turns 
will  result  in  lower  distributed  capacitance. 

Dielectric  losses  generally  are  decreased  by  a  reduction  in  distributed  capacitance. 
However,  the  changes  necessary  to  reduce  the  dielectric  losses  and  distributed  capacitance 
(reducing  the  wire  size,  increasing  the  spacing  between  turns)  finally  begin  to  increase  the 
copper  losses  in  the  conductor  faster  than  the  dielectric  losses  are  decreased.  An  optimum 
design  is  a  compromise  between  all  these  factors,  and  in  a  resonant  circuit  the  distributed 
capacitance  is  frequently  of  lesser  importance  than  the  Q  of  the  inductor;  therefore,  the 
capacitance  is  disregarded,  while  efforts  are  directed  toward  the  attainment  of  minimum 
total  losses.  In  r-f  choke  coil  design,  a  minimum  value  of  distributed  capacitance  is  usually 
desired,  while  a  Q  higher  than  5  or  10  produces  a  negligible  effect.  Therefore,  choke  coils 
are  frequently  wound  with  a  minunum  wire  size,  on  long  slender  forms,  and  with  relatively 
large  spacing  between  turns. 

12.  ELECTRICAL  DESIGN  CONSIDERATIONS 

TYPES  OF  CONDUCTORS.  Copper  wire  is  most  commonly  used  in  inductors.  Copper 
tubing,  or  strips,  are  sometimes  resorted  to  at  frequencies  above  about  50  me,  Litz  wire, 
composed  of  a  number  of  strands  of  fine  enameled  wire  (from  about  No.  38  to  No.  44), 
produces  lower  r-f  resistance  than  a  single  wire  of  the  same  area  of  cross-section.  It  is 
most  effective  in  the  lower-frequency  part  of  the  radio  spectrum,  below  about  2  me. 
Above  this  frequency  the  r-f  currents  appear  to  flow  along  the  outside  of  the  group  of 
conductors  and  the  effectiveness  of  litz  wire  is  not  so  apparent. 

Bare  copper  wire  is  rarely  used  for  inductors.  Tinned  wire  is  found  occasionally  on 
space-wound  solenoids;  silver  plating  is  also  used  occasionally  on  the  heavier  conductors 
normal  for  this  type  of  coil.  Silver  plating  offers  the  advantages  of  high  conductivity  on 
the  surface  of  the  conductor  where  higher-frequency  currents  flow,  soldering  to  the  con- 
ductor is  made  easy,  and  an  attractive  and  fairly  durable  finish  is  produced. 

TYPES  OF  WIRE  INSULATION.  Silk  and  cotton  have  been  the  most  common  mate- 
rials for  wire  insulation  on  air-core  inductors.  Coil  wire  is  made  with  a  spiral  wrapping, 
adding  about  0.002  in.  to  the  diameter  of  the  wire  for  silk  or  about  0.004  in.  for  cotton. 
This  ia  called  a  "serving"  of  silk  or  cotton.  A  double  serving  may  be  used,  with  the  second 
serving  spiraling  in  a  direction  opposite  to  that  of  the  first  serving.  This  is  known  as  double- 


ELECTRICAL  DESIGN   CONSIDERATIONS  3-33 

silk  or  double-cotton  insulation;  it  is  usually  designated  D.S.  or  D.C.  For  instance,  a 
No.  38  bare  wire  with,  double  silk  insulation  could  be  described  as  38  D.S.  Celanese  and 
nylon  are  rapidly  replacing  silk  and  cotton  as  insulation  for  coil  wire.  Braided  fabric 
insulation  is  also  used  to  some  extent. 

An  enamel  coating  is  used,  either  alone  on  wire  for  solenoids,  or  with  one  or  more  servings 
of  silk  or  one  of  the  other  serving  materials  over  it.  The  enamel  adds  about  0.001  in.  to 
the  wire  diameter.  Plastic  coatings  can  be  added  to  conductors,  and  almost  any  required 
outside  diameter  can  be  produced  in  this  manner. 

Litz  wire,  as  it  is  made  in  this  country,  consists  of  three  or  more  strands  of  enameled  wire 
with  one  or  more  servings  of  silk,  cotton,  celanese,  or  nylon,  around  the  group  of  wires. 
The  strands  are  either  twisted  in  a  regular  fashion  as  they  are  wrapped  or  simply  placed 
side  by  side  parallel  to  each  other.  The  twisted  method  produces  a  slightly  higher  Q  under 
some  conditions. 

BEST  COIL  SHAPE.  Formulas  are  available  which  express  the  best  shape  factor  and 
winding  pitch  of  a  single-layer  solenoid  under  idealized  conditions.  In  a  practical  design 
problem  these  formulas  serve  as  a  valuable  guide. 

It  has  been  shown  that  with  a  given  length  of  wire,  wound  with  a  given  pitch,  the  single- 
layer  coil  which  has  the  maximum  inductance  value  is  so  shaped  that  the  ratio 
Diameter  -f-  Width  of  winding  =  2.46,  approximately. 

Brooks  has  determined  that  there  is  a  most  efficient  multilayer  coil  form  to  produce 
the  maximum  inductance  with  a  given  length  of  conductor.  This  most  efficient  inductor 
was  produced  as  a  compact  multilayer  cylindrical  coil  with  a  mean  diameter  2.95  times 
the  side  of  the  square  cross-section.  Other  proportions  varying  somewhat  from  the  op- 
timum affect  the  inductance  only  slightly.  It  has  been  determined  that,  when  the  ratio 
of  mean  diameter  to  side  of  square  cross-section  is  2.80,  the  resulting  inductance  is  only 
0.04  per  cent  less  than  the  maximum  value.  It  is  generally  convenient  and  within  limits 
of  accuracy  to  consider  the  optimum  form  as  having  the  dimensions:  diameter  equal  to 
3  times  width,  and  width  equal  height  of  winding. 

Precision  design  can  be  effected  only  when  the  inductor  is  solely  for  low  frequencies. 
The  problem  then  of  constructing  an  inductor  of  definite  inductance  for  radio  frequencies 
resolves  mainly  into  the  problem  of  minimizing  the  resistance  and  distributed  capacitance. 
A  coil  is  designed  for  a  certain  range  of  frequencies,  and  generally  an  attempt  is  made  to 
construct  a  coil  with  a  uniformly  high  value  of  Q  in  this  range.  In  consideration  of  these 
and  other  requirements,  a  coil  design  will  generally  depart  from  the  optimum  proportions 
indicated  above. 

It  will  be  found  that  a  practical  coil  is,  in  general,  somewhat  elongated  in  the  direction 
in  which  distributed  capacitance  will  be  minimized.  For  instance,  a  practical  solenoid 
is  usually  longer  in  proportion  than  is  indicated  by  the  formula  above;  a  diameter-to- 
length  ratio  of  slightly  over  unity  is  common.  The  universal  winding  is  frequently  elon- 
gated in  the  radial  direction.  A  diameter-to-length  ratio  of  about  0.7  was  found  to  produce 
the  maximum  Q  over  the  broadcast  band  for  a  progressive  universal  winding  on  a  0.5-in.- 
diameter  Bakelite  form. 

Dielectric  and  eddy-current  losses,  which  usually  are  neglected  in  the  computation  of 
"best"  coil  shapes,  appear  to  be  primarily  responsible  for  the  noticeable  discrepancies 
between  calculated  and  measured  data. 

SOLENOID  WINDINGS.  Single-layer  solenoids  are  sometimes  wound  with  each  turn 
touching  the  preceding  turn;  this  results  in  fairly  high  values  of  distributed  capacitance 
and  eddy-current  and  dielectric  losses.  The  length  of  the  winding,  and  consequently  its 
inductance,  vary  considerably  from  coil  to  coil  in  this  type  of  winding  because  of  variations 
in  wire  size  (unless  the  wire  is  selected  to  closer  than  the  usual  limits  for  its  gauge)  and 
because  of  the  failure  of  each  turn  to  lie  snugly  against  the  next  throughout  its  length. 
Therefore,  when  the  inductance  tolerance  of  a  coil  is  closer  than  a  few  per  cent,  it  is  cus- 
tomary to  "spin"  a  few  turns — about  5  to  10  per  cent  of  the  total  on  the  end  of  the  coil. 
These  turns  are  wound  with  the  same  spacing  as  the  main  portion  of  the  winding,  but  the 
group  of  wires  in  the  spun  portion  is  spaced  about  1/8  i*1-  away  from  the  balance  of  the  coil. 
These  "spun"  turns  can  be  moved  on  the  coil  form  nearer  to,  or  away  from,  the  balance 
of  the  winding. 

It  is  possible  to  wind  a  solenoid  to  very  accurate  length  and  inductance  limits  (on  a  form 
of  accurate  diameter)  on  a  winding  machine  or  lathe  adjusted  to  produce  the  desired 
pitch  per  turn.  This  method  allows  uninsulated  wire  to  be  wound  in  a  solenoid,  if  care  is 
taken  to  secure  the  turns  properly  on  the  form  to  prevent  slippage  after  the  coil  is  wound. 
Depending  upon  the  design  requirements,  the  pitch  may  be  chosen  to  give  the  minimum 
spacing  necessary  to  assure  mechanical  and  electrical  uniformity  with  maximum  variations 
in  wire  and  insulation  dimensions,  or  the  spacing  may  be  chosen  for  optimum  electrical 
performance. 


3-34  RESISTORS,   INDUCTORS,   CAPACITORS 

Solenoids  are  sometimes  wound  in  a  screw-thread  groove  in  the  coil  form,  but  unless  a 
molded  form  is  used  it  is  difficult  to  maintain  an  accurate  effective  diameter  of  the  groove 
and  inductance  variations  result.  Also,  winding  the  conductor  in  a  groove  may  reduce 
somewhat  the  Q  of  a  very  efficient  inductor  because  of  increased  dielectric  losses  in  the 
material  in  the  immediate  vicinity  of  the  conductor.  A  slight  increase  in  distributed 
capacitance  will  also  result. 

A  simple  manner  in  which  to  wind  a  single-layer  solenoid  is  to  choose  insulation  of  the 
proper  thickness  to  space  the  turns  properly  when  the  insulated  wire  is  close  wound. 

THE  UNIVERSAL  WINDING.  This  type  of  inductor  is  wound  in  single  or  multiple 
sections  ranging  in  widths  from  about  1/16  to  1/2  in.  On  a  simple  type  of  winding  machine, 
the  width  of  the  coil  is  controlled  by  a  cam  which  oscillates  the  wire  guide  back  and  forth 
in  a  linear  fashion  on  the  periphery  of  the  form  or  on  the  next  lower  layer  of  the  same 
winding.  The  cam  is  geared  to  the  main  shaft  of  the  winding  machine.  The  main  shaft 
holds  the  coil  form  and  rotates  the  coil  as  it  is  wound.  The  number  of  teeth  in  the  gear 
on  the  main  shaft  (or  driven  by  the  main  shaft  through  a  1/1  gear  ratio)  over  the  number 
of  teeth  in  the  gear  on  the  camshaft  is  called  the  gear  ratio.  The  wire  is  wound  on  the 
form  at  an  angle  to  the  side  of  the  coil  form  (winding  angle)  determined  by  the  width  of 
the  cam,  the  diameter  of  the  coil,  and  the  gear  ratio.  Simon  has  stated  that  a  practical 
limit  of  this  winding  angle  is  about  12  deg  maximum.  Above  this  value,  the  wire  may 
slip  on  the  coil  form  and  a  poor  winding  will  result.  The  winding  angle  becomes  smaller 
as  the  coil  builds  up,  being  approximately  inversely  proportional  to  the  diameter  of  the 
winding  at  any  point;  when  this  angle  is  reduced  to  about  6  deg  the  turns  cross  each 
other  at  an  angle  which  is  too  small  and  tend  to  align  themselves  in  the  spaces  between 
adjacent  turns  on  the  previous  layer.  The  coil  will  not  build  up  properly  after  this  point 
is  reached. 

When  a  coil  must  be  wound  up  to  an  outside  diameter  about  equal  to,  or  greater  than, 
twice  its  inside  diameter,  it  is  frequently  necessary  to  tolerate  some  slippage  of  the  wire 
on  the  form  at  the  start  of  the  winding.  The  winding  problem  is,  of  course,  greatly  facil- 
itated when  the  designer  restricts  the  height  of  the  winding  to  a  point  within  practical  limits. 

The  winding  angle  is  proportional  to  the  cam  width  and  also  to  the  gear  ratio,  since  the 
gears  drive  the  cam  at  a  rate  depending  upon  their  relative  number  of  teeth. 

If  gf,  the  pattern  gear  ratio,  is  the  fraction  q'/s',  where  both  qr  and  s'  are  small  whole 
numbers,  a  simple  and  practical  winding  pattern  should  result.  The  number  of  cam 
cycles  per  winding  cycles  is  q',  and  the  number  of  planes  cutting  the  periphery  of  the 
winding  where  interlaced  crossing  of  turns  occur  is  (s'  —  1) . 

In  order  to  have  the  winding  pattern  repeat  on  consecutive  winding  cycles  with  the 
required  spacing  between  adjacent  turns,  the  pattern  gear  ratio  may  be  corrected  by  the 
amount  ±g'xw/2cqf,  where  x  is  the  desired  number  of  wire  diameters  between  turns 
(usually  about  1.25),  and  w  is  the  diameter  of  the  wire.  The  plus  sign  produces  a  retro- 
gressive winding;  the  minus  sign  produces  a  progressive  winding.  The  gear  ratio,  g,  may 
be  computed  g  -  g'(l  ±  [0.63^/c^]). 

THE  PROGRESSIVE  UNIVERSAL  WINDING.  The  progressive  universal  winding  is  a 
special  type  of  universal  winding  in  which  the  wire  guide  is  moved  parallel  to  the  axis  of 
the  coil  form  as  the  coil  winds.  The  machine  for  winding  this  type  of  coil  is  usually 
equipped  with  100  *  1  reduction  gears  driving  the  set  of  gears  on  the  rear  of  the  machine 
which  produce  the  progression.  The  pitch  of  the  winding  produced  by  a  1:1  set  of  pro- 
gression gears  is  usually  very  close  to  0.01  in.,  but  not  exactly  so  on  all  machines. 

It  is  necessary  to  correct  the  gear  ratio  (which  would  be  used  for  an  ordinary  universal 
winding)  slightly  in  order  to  allow  for  the  change  in  spacing  which  results  from  the  pro- 
gression. This  can  be  done  by  reducing  g'  by  the  amount  —  0.5gpp/c,  where  gp  is  the 
progression  gear  ratio  and  p  is  the  pitch.  This  factor  is  usually  rather  small. 

The  steps  required  to  determine  the  winding  machine  setup  for  this  type  of  winding 
follow: 

First,  determine  the  number  of  turns  per  inch  by  dividing  the  desired  number  of  turns 
by  the  required  length.  For  a  relatively  wide  cam  and  short  winding,  it  is  desirable  to 
subtract  one  cam  width  from  the  required  length  for  this  calculation  in  order  to  allow  for 
the  tapering  off  of  the  winding  at  its  ends.  At  this  point  it  is  best  to  examine  the  result 
and  determine  the  number  of  layers  of  wire  which  will  be  built  up  radially  on  the  coil. 
If  the  number  is  less  than  about  2.5  a  solenoid  would  be  preferable;  if  it  is  more  than  about 
5  layers,  the  progressive  winding  may  not  build  up  satisfactorily,  and  a  plain  universal 
winding  of  one  or  more  sections  may  be  better. 

The  cam  for  a  progressive  winding  is  usually  as  narrow  as  possible,  since  the  distributed 
capacitance  and  dielectric  losses  increase  with  a  wider  cam.  A  wider  cam,  however,  de- 
creases the  steepness  of  the  slope  of  the  pile  of  wire  upon  which  a  turn  must  be  wound  and, 
consequently,  enables  a  coil  having  more  layers  to  be  wound. 


ELECTRICAL  DESIGN   CONSIDERATION'S  3-35 

The  value  of  g'  for  a  progressive  winding  is  usually  less  than  for  a  universal,  since  it  is 
unnecessary  to  make  provisions  for  winding  the  coil  up  to  a  height  comparable  to  the  form 
diameter.  In  this  case  it  is  desirable  to  choose  q'  and  s',  again  fairly  small  whole  numbers, 
but  values  such  as  9/s  or  7/n,  where  qf  and  s'  are  larger,  produce  a  better  pattern  on  the 
surface  of  the  coil.  Landon  and  Joyner's  "composite"  winding  is  obtainable  when  a  fairly 
complicated  pattern  is  obtained. 

SHIELDING.  The  successful  operation  of  many  of  the  modern  communication  lab- 
oratory devices  and  radio  receiving  sets  depends  upon  the  effectiveness  of  the  shielding 
between  the  various  parts,  and  so,  in  high-gain  amplifiers,  leads,  vacuum  tubes,  trans- 
formers, and  tuning  coils  are  all  shielded.  This  process  generally  requires  the  placing  of  a 
metallic  shielding  container  around  the  individual  parts.  Shielding  is  attempted  against 
both  electric  and  magnetic  fields  and  is  particularly  necessary  in  circuits  carrying  high- 
frequency  currents. 

As  has  been  pointed  out  before,  whenever  material  is  brought  into  the  influence  of  the 
electric  and  magnetic  fields  of  an  inductor  there  is  a  transfer  of  energy  to  that  material. 
Parts  of  the  inductor  are  generally  at  a  potential  higher  than  that  of  the  shield,  which 
is  usually  at  "ground"  potential.  With  this  condition  there  is  added  to  the  distributed 
capacitance  of  the  winding  more  capacitance  to  "ground."  This  further  complicates  the 
calculation  of  the  actual  inductance  of  the  coil  at  high  frequencies. 

Shields  are  generally  constructed  of  non-magnetic  or  magnetic  metals  such  as  iron,  zinc, 
copper,  and  aluminum.  In  the  shield  used  for  the  shielding  of  a  high-frequency  magnetic 
field,  the  efficiency  of  the  screen  depends  upon  the  eddy  currents  produced  in  the  shield. 
The  energy  involved  in  the  circulation  of  these  eddy  currents  is  drawn  from  the  field  of 
the  shielded  inductor.  Magnetic  shielding  is  therefore  always  accompanied  by  an  increase 
in  the  effective  resistance  of  the  shielded  inductor. 

When  the  resistivity  and  thickness  of  the  metal  shield  remain  constant  and  the  fre- 
quency of  the  alternating  electromagnetic  field  varies,  the  shielding  increases  as  the 
frequency  increases,  owing  to  the  increased  flow  of  eddy  currents.  For  a  given  kind  of 
metal  at  any  specified  frequency  the  shielding  efficiency  increases  as  the  shield  thickness 
is  increased.  Under  these  conditions,  a  certain  thickness  of  shield  introduces  a  maximum 
resistance  into  the  shielded  circuit.  The  thickness  of  the  shield  which  gives  maximum 
added  resistance  to  the  shielded  circuit  decreases  as  the  frequency  increases.  If  shielding 
is  to  be  obtained  by  eddy  currents  they  must  be  free  to  flow  as  they  will,  which  requires 
that  there  be  no  imperfect  joints  or  breaks  in  the  shield.  A  short-circuited  coil  may  be 
used  as  a  shield  since  the  current  induced  in  it  by  the  field  will  set  up  an  opposing  field 
and  give  a  zero  local  resultant. 

In  an  "open"-circuit  electrostatic  screen,  no  eddy  currents  will  flow,  and  the  shield 
may  be  used  to  prevent  the  alternating  field  from  reaching  an  impure  dielectric  and  thus 
producing  a  loss.  In  this  application  the  shield  reduces  the  effective  resistance  of  the  elec- 
trical circuit. 

Since  the  eddy  currents  hi  a  shield  set  up  a  magnetic  field  opposing  the  field  of  the 
inductor,  it  is  evident  that  there  will  be  a  reduction  of  the  net  field  surrounding  the  in- 
ductor winding.  There  is  therefore  a  change  in  the  effective  coil  inductance,  which 
results  in  a  decrease  of  the  inductance  value. 

Many  investigators  have  attempted  to  state  quantitatively  the  magnitude  of  the 
screening  effects  on  coil  inductance  and  resistance.  An  idealized  mathematical  solution 
of  the  problem  (given  in  the  Wireless  Engineer]  replaces  the  ordinary  cylindrical  screen 
by  a  spherical  one,  and  the  cylindrical  coil  by  a  dipole  of  the  same  magnetic  moment  placed 
at  the  center  of  the  sphere.  The  development  is  possible  because  it  has  been  found  that 
the  exact  shape  of  the  screening  can  is  not  important,  and  this  permits  the  mathematical 
use  of  a  sphere  with  a  diameter  the  geometric  mean  of  the  three  coordinates  of  the  can. 
The  expression  developed  shows  the  reduction  of  the  effective  inductance  of  the  coil  to 
depend  upon  the  frequency,  material  constants,  the  screen  thickness,  and  a  linear  dimen- 
sion representing  the  can  size 


in  which  Vc  equals  volume  of  the  coil  (winding  section  X  length);  V8  equals  volume  of 
screening  can;  K  equals  a  constant  less  than  1  (K  =  0.7  when  coil  length  —  coil  diam- 
eter); I/o  equals  actual  inductance  of  the  short  solenoid;  and  a  equals  factor  depending 
upon  /,  dimensions,  permeability,  and  resistivity  of  the  screen  can  (a  —  almost  1  for 
non-magnetic  materials)  . 

A  general  interpretation  which  may  be  made  of  this  expression  is  that  the  effect  of  the 
screen  on  the  coil  inductance  varies  inversely  as  thja  diameter  of  the  screen  can  to  the 
cube  power. 


3-36  RESISTORS,  INDUCTORS,   CAPACITORS 

This  conclusion  has  been  stated  by  Hayman,  and  the  results  of  a  simple  expression 
of  the  effect  have  been  accurately  checked  by  experiment.  The  approximate  expression 
which  has  been  given  for  short  coils  is 

(jT}3   _   ££3\ 
ni       j 

where  D  —  screen  diameter  and  d  —  coil  diameter.  As  the  coil  length  approaches  half 
the  can  length  there  is  more  influence  from  the  ends  of  the  can.  This  condition  requires 
the  use  of  a  correcting  factor  which  is  expressed  as 

End  correction  =     1  —  (  —  -  )  (3) 

L          \2Z  can/  J 

The  calculations  with  these  expressions  for  coils  which  do  not  exceed  half  the  can  dimen- 
sions have  checked  experimental  measurements  within  less  than  1  per  cent. 

The  effect  of  the  eddy-current  flow  in  drawing  energy  from  the  inductor  results  in  an 
increase  of  the  effective  resistance  of  the  inductor.  If  skin  effect  in  the  shield  is  negligible 
and  the  eddy  currents  are  uniformly  distributed  through  the  screen^  the  effective  addition 
to  the  coil  resistance  has  been  stated  as 


in  which  A  —  cross-section  area  of  coil,  square  centimeters;  T  —  number  of  coil  turns; 
t  —  thickness  of  screen,  millimeters;  r  =  radius  of  screen  can,  centimeters;  and  p  =  re- 
sistivity of  the  can  material,  ohms  per  cubic  centimeter.  When  skin  effect  in  the  shield 
forces  a  non-uniform  distribution  of  eddy  current  the  above  expression  is  modified  to 
take  the  form 


Rt  =  0.95  X  10~4  !TU2  (5) 

r4 

An  indication  of  the  presence  of  skin  effect  in  the  shield  is  gained  from  the  expression 

'V1' 

*  p 

When  this  factor  is  less  than  5000,  the  skin  effect  is  negligible. 

13.  MECHANICAL  DESIGN  CONSIDERATIONS 

FORM  MATERIALS.  Some  of  the  factors  to  be  considered  in  choosing  a  form  material 
are  its  mechanical  strength,  dielectric  properties,  coefficient  of  thermal  expansion,  machin- 
ability,  moisture  absorption,  power  factor  at  the  operating  frequency,  and  cost.  Sometimes 
the  operating-temperature  requirements  will  not  permit  the  choice  of  an  otherwise  desir- 
able material,  or  the  heat  required  to  solder  leads  to  lugs  on  the  coil  form  may  soften  the 
coil  form  and  loosen  the  solder  lugs. 

It  has  been  found  advisable  to  select  an  extremely  stable  material,  whenever  frequency 
drift  requirements  are  severe,  rather  than  an  unstable  material  with  compensation  else- 
where in  the  circuit  for  inductance  changes.  Glass  coil  forms  are  sometimes  possible  where 
only  a  very  small  thermal  expansion  can  be  permitted.  The  various  kinds  of  glass,  with 
coefficients  of  linear  expansion  of  about  3  to  9  parts  per  million  per  degree  centigrade,  are 
among  the  best  materials  now  available.  Steatite  and  mycalex  also  exhibit  good  tempera- 
ture stability  characteristics. 

While  phenolic  materials  have  been  popular  because  of  their  adaptability  for  use  as 
form  materials  and  their  relatively  low  cost,  their  coefficient  of  linear  expansion  is  only 
fair,  being  about  30  parts  per  million  per  degree  centigrade. 

The  coefficient  of  thermal  expansion  of  materials  which  are  not  homogeneous  (such  as 
commercial  laminated  Bakelite  tubing)  frequently  has  different  values  for  the  radial  and 
axial  dimensions.  Average  values  of  these  two  coefficients  can  generally  be  furnished  by 
the  manufacturer  or  may  be  measured  directly. 

The  coil  form  is  usually  much  more  rugged  than  the  winding,  and  therefore  the  tempera-* 
ture  instability  of  a  coil  is  usually  due  to  its  form.  Copper  wire  expands  at  about  the  same 
rate  as  Bakelite  under  varying  temperature  conditions.  However,  copper  wire  can  be 
wound  under  tension  on  one  of  the  more  stable  form  materials  and  a  stable  coil  will  result; 
the  form,  material  must  be  sufficiently  stronger  than  the  wire  so  that  the  form  is  not  dis- 
torted by  the  pressure  of  the  wire. 


MECHANICAL  DESIGN  CONSIDERATIONS 


3-37 


Absorption  of  water  by  the  coil  form  may  result  in  loss  of  mechanical  strength,  a  change 
in  the  distributed  capacitance  of  the  inductor,  and  consequent  detuning  of  the  circuit, 
and  may  produce  conditions  favorable  to  electrolysis  or  fungus  growth.  The  various 
methods  of  treating  coils,  such  as  impregnating,  varnishing,  or  the  application  of  fungi- 
cides, may  delay  these  harmful  results,  but  it  appears  that  the  most  certain  way  to  avoid 
them  is  to  choose  materials  which  will  not  absorb  water,  if  such  materials  are  available 
and  fulfill  the  other  design  requirements. 

A  partial  list  of  the  more  important  properties  of  some  of  the  more  frequently  used  form 
materials  is  given  herewith. 

Typical  Properties  of  Form  Materials  * 


Form 
Material 

Dielectric 
Constant 

Power 
Factor, 
%  at  1  me 

Maximum 
Temperature, 
deg  cent 

Coefficient  f 
of  Linear 
Thermal 
Expansion 

Water 
Absorption, 
%  in  24  hr 

Machin- 
ability 

Bakelite,  molded.  . 
paper  base  

6.0 
5.5 

4.0 
4.0 

120 
120 

30.0 
30.0 

0.2 
0  2 

Fair 
Good 

Glass      

6.0 

0.4 

Over  500 

8.5 

0 

Very  poor 

Pyrex              .... 

4.5 

0.2 

500 

3  5 

o 

Magnesium  silicate 

6.0 
7  0 

0.3 
0  3 

1000 
300 

7.0 
8  5 

0.02 
0  04 

Very  poor 

Polystyrene 

2  6 

0.03 

75 

70  0 

0  01 

Good 

Porcelain  .  .        ... 

6.5 

0.7 

1000 

4.0 

0  5 

Very  poor 

Rubber,  hard  

3.0 

1.0 

65 

75.0 

0.01 

Fair 

*  Values  given  are  subject  to  considerable  variation, 
f  In  parts  per  million  per  degree  centigrade. 

IMPREGNATION  OF  INDUCTORS.  Coil  impregnation  serves  two  principal  purposes: 
first,  the  impregnating  material  tends  to  seal  out  moisture;  second,  the  impregnating  mate- 
rial improves  the  mechanical  strength  of  the  winding  and  holds  it  more  firmly  in  place 
on  the  coil  form. 

Either  before  or  during  the  impregnation  of  the  coil,  it  is  necessary  to  drive  out  of  the 
winding  and  form  any  moisture  that  may  be  present.  This  is  accomplished  either  by 
baking  the  coil  at  a  temperature  slightly  higher  than  100  deg  cent  before  impregnating  or 
by  maintaining  the  impregnating  material  at  such  a  temperature  while  the  coil  is  being 
impregnated.  The  maximum  limit  of  the  temperature  used  during  baking  or  impregnating 
is  the  temperature  that  will  damage  some  part  of  the  winding  or  its  form  or  the  impreg- 
nating material. 

There  are  many  types  and  mixtures  of  different  types  of  waxes  which  are  used  to  im- 
impregnate  coils.  Resins  also  are  mixed  with  waxes  to  improve  their  characteristics. 
The  melting  and  softening  temperatures,  and  the  hardness  of  the  wax,  are  determined  by 
the  kinds  and  amounts  of  the  various  kinds  of  waxes  in  the  mixture.  Some  mixtures  be- 
come extremely  brittle  at  fairly  low  temperatures,  which  may  be  a  serious  disadvantage. 

Varnish  of  good  electrical  quality,  or  polystyrene  dissolved  in  a  solvent,  are  sometimes 
used  to  impregnate  coils.  Vacuum  impregnation  (dipping  the  coil  in  the  impregnating 
material  while  it  is  in  a  partial  vacuum)  is  usually  resorted  to  when  varnish  or  some  similar 
material  is  used  to  impregnate  a  coil. 

The  conditions  under  which  the  coil  will  be  used  determine  the  designer's  choice  of  the 
method  and  material  for  impregnating  the  coil.  Very  often  the  economy  of  the  method  is  a 
determining  factor. 

Coils  which  are  designed  for  good  conditions  of  temperature  and  humidity,  such  as  in 
equipment  to  be  used  indoors  in  all  but  the  most  humid  parts  of  the  United  States,  are 
impregnated  usually  in  a  simple  and  inexpensive  way.  For  instance,  many  coils  are  treated 
by  baking  in  an  oven  until  most  of  the  moisture  is  driven  out,  followed  by  immersion  in  a 
wax,  resin,  or  mixture  of  both,  at  a  temperature  slightly  higher  than  the  boiling  point  of 
water  until  all  agitation  of  the  liquid  ceases.  The  coil  is  then  cooled  and  finally  "flash- 
dipped"  to  produce  an  even  coating  of  the  same  or  a  different  impregnant  on  the  outer 
surface.  The  first  (baking)  operation  is  frequently  omitted  for  the  sake  of  economy. 

Where  more  severe  conditions  of  temperature  and  humidity  are  encountered,  some  better 
methods  of  impregnation  and  better  materials  are  required.  Two  such  impregnation 
procedures  are  described  below.  These  methods  were  developed  and  tested  by  the  Hazel- 
tine  Corporation  for  use  on  equipment  for  the  Army  and  Navy  during  1945.  The  two 
processes  are: 

Process  1.  Q-Max  A-27  diluted  1:1  with  Toluene,  applied  by  dipping  and  baking; 
recommended  for  Silicone-varnished  Fibreglas-served  wire. 


3-38  RESISTORS,   INDUCTORS,   CAPACITORS 

(a)  Bake  the  coil  (on  its  form)  for  1  hr  at  110-120  deg  cent. 

(6)   While  still  hot,  immerse  the  coil  until  bubbling  ceases  in  a  solution  consisting  of: 

1  part  Q-Max  Lacquer  A-27 

1  part  Toluene  ("technical  grade") 

(c)  Drain  and  air  dry  for  1  hr. 

(d)  Bake  for  4  hr  at  140  deg  cent. 

(e)  Apply  two  more  coats  per  (&) ,  (c) ,  and  (d)  above,  redipping  immediately  after  (d) . 
Both  processes  described  here  require  the  use  of  combustible  materials.     Adequate 

ventilation  must  be  provided  and  safety  rules  regarding  fire  hazards  observed. 

Process  2.  Styrene  Monomer  N-100  and  Q-Max  A-27  diluted  1 : 1  with  Toluene,  applied 
by  dipping  and  baking.  Recommended  for  use  on  Fibreglas-  or  silk-served  wire. 

(a)  Bake  the  coil  (on  its  form)  for  2  hr  at  110  deg  cent. 

(6)   While  still  hot,  immerse  the  coil  in  styrene  monomer  for  20  min. 

(c)  Air  dry  the  coil  until  dripping  ceases. 

(d)  Bake  for  24  hr  at  125  deg  cent. 

(e)  Redip  in  styrene  monomer  for  20  min. 
(/)  Repeat  (c)  and  (d). 

(g)   Apply  a  thin  coat  of  the  Q-Max  and  Toluene  solution. 

Coils  impregnated  according  to  the  two  processes  given  above  should  withstand  tem- 
peratures ranging  from  —  65  to  +85  deg  cent  with  the  relative  humidity  as  high  as  95  per 
cent  at  the  highest  temperature. 

THE  SPECIFICATION  OF  INDUCTORS.  The  materials  and  the  method  of  construc- 
tion of  inductors  can  be  shown  readily  by  means  of  drawings  and  written  specifications. 
However,  it  is  sometimes  difficult  to  specify  the  performance  requirements  of  an  inductor 
in  an  exact  fashion.  This  is  due  to  a  lack  of  standardized  measuring  equipment  in  the 
industry  capable  of  separating  the  effects  of  the  distributed  capacitance,  inductance,  and 
Q  of  the  inductor  in  a  practical  and  accurately  measurable  fashion,  especially  on  the  higher- 
frequency  coils.  It  is  possible,  however,  to  compare  one  inductor  to  another  with  a  higher 
degree  of  precision.  Therefore,  it  has  become  a  fairly  widely  accepted  practice  for  the 
designer  to  adjust  accurately  one  complete  set  of  coils,  which  become  the  master  standards. 
From  these  master  standards,  as  many  secondary  standards  as  are  required  can  be  made 
and  distributed  to  the  coil  adjusting  and  testing  points.  All  performance  specifications 
requiring  very  close  tolerances  are  referred  to  these  standards.  If  the  standards  are  care- 
fully prepared,  stored,  and  handled,  the  variations  in  their  performance  over  a  period  of 
time  due  to  aging,  etc.,  are  minimized.  It  is  desirable  to  obtain  measurements,  on  the 
most  stable  equipment  available,  of  all  possible  performance  data  of  the  standards  so  that 
the  master  standards  themselves  can  be  rechecked.  If  both  the  coil  manufacturer  and 
designer  can  measure  the  inductor,  with  similar  equipment  and  by  similar  methods,  the 
data  and  the  inductors  measured  can  be  exchanged  and  the  test  equipment  calibrated  alike 
in  both  places.  Even  if  this  can  be  done,  in  view  of  the  difficulty  involved  in  obtaining 
standardized  test  equipment  and  conditions  it  is  generally  desirable  to  use  the  master 
standard  as  the  basis  of  the  specification  of  inductor  performance. 

14.  INDUCTOR  DESIGN  FORMULAS 

The  formulas  that  follow  have  been  found  to  be  generally  useful  in  the  design  of  induc- 
tors. The  inductance  formulas  are  less  accurate  than  those  given  in  the  Bureau  of  Stand- 
ards Circular  C74,  They  do,  however,  produce  a  result  with  approximately  the  same  accu- 
racy with  which  the  usual  range  of  radio-frequency  inductors  can  be  wound,  using  materials 
which  are  not  selected  to  closer  than  normal  limits.  Their  principal  advantage  is  their 
simplicity. 

The  dimensions  and  symbols  used  in  these  formulas  follow: 

L  =  inductance  of  each  section  of  a  winding  in  microhenries. 

d  —  form  diameter  in  inches. 

"b  —  length  of  winding  in  inches. 

c  —  throw  of  cam  used  to  wind  a  universal  section  in  inches. 
w  =  outside  diameter  of  wire,  including  covering  insulation,  in  inches. 

t  =  number  of  turns. 

C0  =  distributed  capacitance  in  micro-microfarads, 
p'  =  pitch  of  winding  in  inches. 

h  —  height  of  winding  above  coil  form  in  inches. 

a  =  mean  diameter  of  winding  in  inches. 


INDUCTOR  DESIGN  FORMULAS  3-39 

£,t  =  total  inductance  of  a  multisection  universal  winding  in  microhenries. 
gp  =  progression  gear  ratio. 
y  =5  number  of  sections  in  winding. 
g  =  gear  ratio. 
g'  =  pattern  gear  ratio. 

qf  —  small  whole  number,  numerator  of  fraction  gf. 
sf  =  small  whole  number,  denominator  of  fraction  gf. 
z  =  number  of  strands  in  litz  wire. 

La  =  apparent  inductance  in  microhenries  of  coil  with  distributed  capacitance. 
o>  =  2ir  times  frequency  in  megacycles  per  second. 
p  —  pitch  produced  by  winding  machine  when  1/1  progression  gear  ratio  is  used. 

SOLENOID  AND  PROGRESSIVE  UNIVERSAL  WINDING.    Inductance. 


I8d  +  406 

Accurate  to  within  about  1  per  cent  for  solenoids  with  b  >  QAd. 
Turns. 

t  =  VL(lSd  +  406)  (7) 

d 
Distributed  Capacitance. 

C°  =  cosh^  p'/w  (8) 

for  single  layer  solenoids  only,  where  w  is  the  diameter  of  the  bare  wire.  The  dielectric 
constant  of  the  coil  form,  wire  insulation,  and  impregnating  material  will  increase  the 
distributed  capacitance  above  the  calculated  value. 

For  progressive  universal  windings,  the  distributed  capacitance  is  the  minimum  for  the 
smallest  cam  throw;  it  is  approximately  proportional  to  the  coil  diameter. 

EACH  SECTION  OF  A  UNIVERSAL  WINDING.    Inductance. 


~  3d  +  9c  +  10^ 

Accuracy  to  within  about  1  per  cent  when  the  three  terms  in  the  denominator  are  about 
equal. 
Turns. 

t  =  log-1  (1.08  -  QA7d  +  0.16c  +  0.5  log  L)  (10) 

to  within  5  per  cent  approximately,  for  windings  having  more  than  about  100  turns.    This 
is  an  empirically  derived  formula.    If  more  accurate  results  are  required,  the  result  may 
be  substituted  in  eq.  (6)  above,  and  a  suitable  correction  obtained. 
Height  of  Winding. 

(11) 


for  wire  which  does  not  flatten  appreciably  during  winding,  and  with  1.25-wire-diameter 
spacing.  For  litz  wire,  flattening  of  the  insulation  may  reduce  the  calculated  height  by 
about  5  to  10  per  cent. 

Distributed  Capacitance.  The  distributed  capacitance  of  a  universal  winding  is  ex- 
tremely  variable;  it  depends  upon  many  factors,  including  the  pattern  gear  ratio  (g')»  the 
spacing  between  conductors,  etc.  It  is  approximately  proportional  to  the  throw  of  the 
cam  and  to  the  form  diameter.  It  is  fairly  independent  of  the  number  of  turns  on  the 
winding. 

MULTISECTION  UNIVERSAL  WINDING.    Inductance. 

1/2  =  L&y  -  1)  (12) 

approximately,  for  each  section  with  uniform  spacing  between  sections  about  equal  to 
the  width  of  the  section. 

Distributed  Capacitance.  Slightly  greater  than  the  resultant  of  the  distributed  capaci- 
tances of  the  individual  sections  in  series. 

SPIRAL  WINDING.    Inductance. 

(d  +  7W 


16d  +  44A 
where  h  is  the  difference  between  outside  and  inside  diameters. 


(13) 


3-40  RESISTORS,   INDUCTORS,   CAPACITORS 

MEASUREMENT  FORMULAS.    Apparent  Inductance. 

La  =  1  -  rfLCo  (14) 

Distributed  Capacitance. 

Co  =  <^  (15) 

where  Ci  and  C*  are  the  capacitances  required  to  resonate  the  inductor  at  frequencies  / 
and  2/f  respectively.  When  C\  and  C%  are  large,  the  value  of  C0  obtained  will  be  somewhat 
larger  than  that  computed  from  the  actual  self-resonance  of  the  coil. 

FIGURE  OF  MERIT.  The  Q  of  an  inductor  is  usually  measured  directly  on  equipment 
designed  for  that  purpose.  If  such  equipment  is  not  available,  a  voltage  can  be  induced 
in  the  inductor  in  a  resonant  circuit,  with  suitable  precautions  taken  to  avoid  additional 
loading,  and  the  resonant  band  width  measured  at  the  half-power  point.  The  ratio  of 
the  test  frequency  to  the  distance  between  the  half-power  points  is  equal  to  the  Q  of  the 
inductor. 

UNIVERSAL  WINDING  GEAR  RATIO  CALCULATIONS 

Minimum  cam  width  =  Approximately  3w 
Cam  Cycles  per  "Winding  Cycle. 


for  forms  of  average  smoothness;  this  was  expressed  first  by  Simon  in  terms  of  cross-overs 
(one-half  cam  cycle)  per  winding  cycle: 

*  -  p  -  5  (17> 

for  very  smooth  forms.    Choose  a  small  whole  number  for  q'  and  s',  with 

Gear  Ratio. 

/         n  fta*n\ 

(19) 

Set  the  calculated  value  of  g  on  the  C  or  D  scale  of  a  10-in.  slide  rule,  and  locate  the  coinci- 
dence of  two  whole-number  lines  within  the  range  of  available  gears  on  the  C  and  D  scales. 
These  two  coincident  line  numbers  can  be  used  as  the  number  of  teeth  in  the  gears. 

PROGRESSIVE  UNIVERSAL  WINDING  GEAR  RATIO  CALCULATION.    To  deter- 
mine whether  progressive  universal  should  be  used,  if 

—  <  1  (20) 

a  solenoid  should  be  used,  according  to  Landon  and  Joyner.  To  determine  minimum  cam 
throw: 

c  >  —  (21) 

gpP 

which  is  Landon  and  Joyner's  formula  in  the  terms  used  here.  To  determine  cam.  cycles 
per  winding  cycle: 

g'  =  — -  =  i  (22) 

Choose  fairly  small  whole  numbers  for  qf  and  s';  the  more  complicated  patterns  result 
when  sf  is  made  fairly  large,  and  a  proper  choice  of  qf  and  sf  can  be  made  so  that  the  wire 
is  adjacent  to  a  preceding  turn  in  the  pattern  on  both  forward  and  backward  strokes  of 
the  cam,  producing  Landon  and  Joyner's  "composite"  winding.  To  determine  progression 
gear  ratio: 

gp  =  j-  (23) 

To  determine  gear  ratio: 

/  0  Rff^n\    /  n  fi2in\ 

(24) 
This  is  essentially  the  original  formula  published  by  Landon  and  Joyner,  in  the  form  sug- 


BIBLIOGRAPHY  3-41 

gested  by  Simon.  Use  the  minus  sign  after  gf  when  the  winding  forms  a  right-hand  screw 
thread.  If  a  left-hand  thread  is  formed,  a  plus  sign  should  be  used.  The  value  of  g  may 
be  computed  in  the  same  manner  as  for  a  universal  winding. 

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Grover,  F.  W.,  Additions  to  the  Formulas  for  the  Calculation  of  Mutual  and  Self  Inductance,  Bureau 

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Pollack,  D.,  and  Hartley,  M.,  Short  Wave  Coil  Characteristics,  Research  Labs.,  Columbia  University 

(1935).     -    8 

Simon,  A.  W.,  Winding  the  Universal  Coil,  Electronics,  Vol.  9,  22  (October  1936). 
Joyner,  A.  A.,  and  Landon,  V.  D.,  Theory  and  Design  of  Progressive  Universal  Coils,  Communications, 

Vol.  18,  5  (September  1938). 

Hershey,  L.  M.,  The  Design  of  the  Universal  Winding,  Proc.  I.R.E.,  Vol.  29,  442  (August  1941). 
Simon,  A.  W.,  On  the  Winding  of  the  Universal  Coil,  Proc.  I.R.E.,  Vol.  33,  35  (January  1945). 
Simon,  A.  W.,  On  the  Theory  of  the  Progressive  Universal  Winding,  Proc.  I.R.E.,  Vol.  33,  868  (Decem- 
ber 1945). 

Terman,  F.  E,,  Some  Possibilities  for  Low-loss  Coils,  Proc.  I.R.E.,  Vol.  23,  1069  (September  1935). 
Reber,  G.,  Optimum  Design  of  Toroidal  Inductances,  Proc.  I.R.E.,  Vol.  23,  1056  (September  1935). 
Pollack,  D.,  The  Design  of  Inductances  for  Frequencies  between  4  and  25  megacycles,  RCA  Review, 

Vol.  II,  184  (October  1937),  and  Electrical  Engineering,  September,  1937. 


3-42 


RESISTORS,  INDUCTORS,   CAPACITORS 


FERROUS-CORED  INDUCTORS 

By  A.  J.  Rohner 

The  utility  of  ferromagnetic  cores  in  coils  lies  in  the  fact  that  such  cores  have  a  higher 
magnetic  permeability  than  air.  This  permeability  may  be  anywhere  between  2  and 
100,000.  The  use  of  a  ferrous  core  may  have  the  following  beneficial  effects: 

(a)  Increase  of  inductance.  With  a  complete  magnetic  path  of  ferromagnetic  material, 
this  increase  may  be  several  thousand  times  the  air-core  value  of  inductance. 

(6)  Increase  of  Q.  This  results  from  the  increased  inductance,  if  the  increase  of  loss 
due  to  core  loss  is  not  greater  than  the  increase  of  inductance. 

(c)  Magnetic  shielding.     The  magnetic  field  of  the  coil  is  constrained  to  follow  to  a 
large  extent  the  path  of  the  high-permeability  core. 

(d)  Adjustability  of  inductance.    Movement  of  the  core  in  or  out  of  the  coil,  or  variation 
of  an  air  gap  in  the  core,  gives  mechanical  means  of  adjusting  the  inductance. 

The  limitations  of  ferromagnetic  cores  in  coils  are  due  to  certain  undesirable  qualities  of 
ferromagnetic  materials.  Most  important  of  these  are: 

(a)  Magnetic  saturation,  which  occurs  at  7000  to  15,000  lines  per  square  centimeter. 
These  values  depend  upon  the  kind  of  core  material  and  are  for  a  magnetizing  force  of 
10  oersteds. 

(Z>)  Non-constant  permeability.  Permeability  varies  with  the  direct  current  passing 
through  the  coil,  the  alternating  voltage  impressed  across  the  coil  terminals,  and  other 
factors. 

(c)  Core  loss,  which  is  a  wattage  loss  additional  to  the  copper  loss  of  the  coil;  it  deter- 
mines the  frequency  range  for  which  each  type  of  core  material  may  be  used. 

Core  loss  consists  of  two  distinct  parts,  hysteresis  loss  and  eddy-current  loss.  See 
Section  2,  "Magnetic  Materials,"  Spooner,  "Properties  and  Testing  of  Magnetic  Mate- 
rials," or  "Magnetic  Circuits  and  Transformers"  by  staff  of  M.I.T.  Hysteresis  loss  is  a 
magnetic  effect  due  to  the  magnetizing  and  demagnetizing  of  the  core  and  is  proportional 
to  the  frequency.  Hysteresis  loss  can  be  reduced  by  using  core  material  that  is  easily 
magnetized  and  demagnetized,  i.e.,  a  "magnetically  soft"  material.  Eddy-current  loss 
is  an  electrical  effect  due  to  induced  currents  within  the  core  material  and  is  proportional 
to  the  square  of  the  frequency.  Eddy-current  loss  can  be  reduced  by  using  core  material 
of  high  electrical  resistivity,  and  by  laminating  or  powdering  the  core. 

At  the  higher  frequencies,  eddy-current  loss  becomes  predominant  and  a  finer  subdivision 
of  the  core  material  is  necessary.  ^ 

Frequency  Ranges.  At  frequencies  below  about  4  kc,  the  core  usually  consists  of  thin 
sheets,  either  in  the  form  of  fiat  plates  or  "laminations,"  as  thin  as  0.003  in.,  or  in  the  form 
of  ribbon,  which  can  be  made  as  thin  as  0.001  in.  Above  20  kc,  the  core  usually  consists 
of  powdered  material,  the  grains  of  which  may  average  as  little  as  0.00012  in.  in  diameter. 
Between  4  and  20  kc,  either  sheet  material  or  powdered  material  may  be  suitable.  How- 
ever, thin  ribbon  cores  are  useful  up  into  the  low-  and  medium-radiofrequency  bands, 
while  powdered  cores,  of  larger  grain  size,  are  useful  down  to  1  kc  or  lower. 


15.  LOW-FREQUENCY,  SHEET-CORE  INDUCTORS 


(a) 

Scrap  Less  El 


Strapless 


(6) 

Strapless  EE 


FIG.  1. 


W 

Strapless 


Ribbon  core 
Laminations  and  Ribbon  Core 


CORE  CONSTRUCTION. 

Typical  lamination  shapes, 
and  a  ribbon  core,  are  shown 
in  Fig.  1.  In  general,  all 
sheet-material  cores  form  a 
complete  magnetic  path 
around  the  coil,  except  for 
small  air  gaps  which  may  be 
purposely  introduced,  and  the 
cross-section  of  this  magnetic 
path  is  essentially  uniform 
throughout  the  length  of  the 
magnetic  circuit. 

Assembly  of  laminated 
cores  is  illustrated  in  Fig.  2. 
A  butt  joint  is  actually  a 
small  air  gap,  because  of  oxide 
on  the  ends  of  the  lamina- 


LOW-FREQUENCY,  SHEET-CORE  INDUCTORS 


3-43 


tions,  non-squareness  of  the  ends,  and  imperfect  meeting  of  the  joint.  This  butt-joint 
gap  may  be  from  0.0005  to  0.002  in.  long.  A  value  of  0.0015  in.  per  gap  may  be  assumed 
for  average  design  purposes.  When  there  is  an  air  gap,  the  magnetic  flux  usually  must 
cross  this  separation  twice.  Thus,  for  example,  if  the  air-gap  spacer  is  0.010  in.  thick, 
the  total  length  of  air  gap  in  the  core,  including  the  butt-joint  gaps,  is  0.023  in. 

Usually,  ribbon  cores  are  cut  in  two  after  winding,  forming  two  C-shaped  pieces,  and 
the  ends  of  each  C-piece  are  ground  flat.    When  the  two  pieces  are  placed  together,  after 
adding  the  coil,  the  ends  meet  in  tight  butt  joints, 
each  joint  being  about  0.0005  in.  long,  so  that  the  core      ^^H^^        Cross 
as  a  whole  approaches  a  continuous  ferromagnetic    *^^s§§^s.    section 
path. 

Stacking  Factor.  Because  of  oxide  or  other  insula- 
tion on  the  sheet  material,  non-flatness  of  sheets,  and 
stamping  burrs  on  laminations,  the  magnetically  use- 
ful cross-section  of  a  sheet  core  is  never  100  per  cent 
of  the  measured  cross-section.  The  ratio  of  the  two 
is  called  the  "stacking  factor."  Values  that  may  be 
used  as  a  guide  are  given  in  Table  1.  Interleaved 


Mean  length  of 
magnetic  path 


Table  1.     Stacking  Factors 


Laminated  Cores 

Ribbon  Cores 

19-mil  
1  4-mil    
6-mil  
3-mil  

0.94 
0.92 
0.83 
0.71 

14-mil  
5-mil  
3-mil  
2-mil  

0.65 
0.91 
0.86 
0.80 

Air-gap 


Materials  most  commonly  used  in  sheet-material 
cores  are:  Button!  Air  gap 

(a)  Silicon  steel,  especially  the  better  grades,  having  FIG.  2.  Assembly  of  Laminated  Cores 
silicon  content  from  2.5  to  4.75  per  cent. 

(6)   Grain-oriented  silicon  steel.    Hipersil  and  Silectron  are  trade  names  for  this  material. 

(c)  Nickel-iron  alloys,  of  approximately  50-50  composition,  variously  known  as  Nicaloi, 
Hipernik,  4750,  or  49- Alloy. 

(d)  Permalloy,  an  alloy  of  about  80  per  cent  nickel  with  iron.     Hymu  is  a  similar 
material. 

(e)  Mumetal,  similar  to  permalloy,  but  with  5  per  cent  copper  added. 

For  more  detailed  description  of  sheet-core  materials,  see  Section  2,  Magnetic  Materials, 
Spooner,  Chapter  IV;  Elmen,  "Magnetic  Alloys  of  Iron,  Nickel,  and  Cobalt,"  /.  Franklin 
Inst.,  May  1929;  Alleghany  Ludlum  Bulletins  EM-11  and  EM-12,  and  then-  Magnetic 
Core  Materials  Practice;  Follansbee,  Electrical  Sheet  Handbook,  Magnetic  Metals  High 
Permeability  Alloys;  and  Westinghouse,  Metals  and  Alloys. 

Thickness  of  Sheets.  For  inductors  operating  at  25  to  120  cycles,  25-mil-thick  (U.S.S. 
gage  24)  and  19-mil-thick  (26-gage)  laminations  are  useful.  Sheet  of  14-mil  thickness 
(29  gage)  is  widely  used,  both  for  laminations  and  ribbon,  for  applications  from  60  cycles 
to  the  middle  audiofrequencies.  Below  14  mils,  sheet  material  can  be  obtained  in  almost 
any  thickness  down  to  1  mil.  However,  stamped  laminations,  and  preformed  ribbon  cores, 
of  these  thinner  sheets  became  available  largely  as  a  result  of  war  needs  and  are  not  yet 

standardized.  Laminations  can  be  purchased,  of  7- 
mil  silicon  steel,  6-mil  49-Alloy  and  Hymu,  4-mil  4750, 
and  3-mil  4750,  in  a  large  variety  of  sizes.  Hipersil 
ribbon  cores  are  obtainable  in  14-,  5-,  3-,  2-,  and  1-mil 
sheet  thickness. 

COIL  CONSTRUCTION.  Low-frequency  induc- 
tors usually  have  multilayer  coils,  with  insulation 
between  layers.  See  Fig.  3.  The  coil  is  wound  upon 
a  rectangular  spool  of  spirally  wrapped  paper  or  fiber, 
30  or  40  mils  in  total  thickness,  slightly  larger  in  inside 
dimensions  than  the  core  over  which  it  is  to  be  placed, 
and  slightly  shorter  than  the  core  window.  These 
clearances  may  be  */32  or  Vie  in.  Wire  is  usually  solid 
copper,  with  enamel  insulation.  The  length  of  the 
winding,  or  "wire  traverse,"  is  less  than  the  length  of  the  spool,  to  allow  a  "margin" 
of  3/32  to  3/ie  in.  at  either  end.  Over  each  layer  of  wire  is  placed  one  turn  of  insulation, 
the  same  width  as  the  length  of  the  spool,  which  forms  a  smooth  support  for  the  next 
layer.  Kraft  paper,  having  a  thickness  about  Vs  of  the  wire  diameter,  is  a  very  satis- 


Layer  Insul; 


Spool. 


artfn 


Start  of  cofl 


Start  of  second  layer 

FIG.  3.     Layer-wound  Coil 


3-44  RESISTORS,   INDUCTORS;   CAPACITORS 

factory  material  for  layer  insulation.  Glassine  paper,  of  about  the  same  thickness,  is 
often  used  with  wire  of  24-gage  (A.W.G.)  or  smaller.  Layer-wound  coils,  of  this  con- 
struction, require  no  end  boards  to  hold  the  wires  in  place. 

Coils  without  layer  insulation  (random-wound)  may  also  be  used,  allowing  about  50 
per  cent  more  turns  in  a  given  space.  There  are  several  disadvantages  to  this  type  of 
winding,  however.  The  wire  must  have  more  than  ordinary  enamel  insulation  to  prevent 
shorted  turns,  which  reduces  the  space  advantage  somewhat.  End  boards,  or  tape,  are 
necessary  to  hold  the  wires  in  place.  Multiple  winding  cannot  be  used. 

After  the  coil  is  wound,  flexible  lead  wires  are  attached  if  the  wire  of  the  coil  is  smaller 
than  20  gage.  Then  the  coil,  or  the  core  and  coil  together,  are  impregnated  with  a  varnish. 
wax,  or  asphaltic  compound  to  exclude  moisture  and  air  and  to  strengthen  the  coil  mechan- 
ically. See  Belden  Handbook  12,  Anaconda  "Magnet  Wire  and  Coils,"  or  Inca  Bulletin  3. 

Inductors  having  a  three-legged  core,  Fig.  1  (a)  ,  (6)  ,  (c)  ,  have  a  single  coil,  placed  on  the 
middle  leg  of  the  core.  With  two-legged  cores,  Fig.  l(d),  (e),  (/),  two  coils  are  sometimes 
used,  one  on  each  leg,  the  two  coils  being  connected  in  series  or  in  parallel.  The  use  of 
two  coils  results  in  lower  resistance  and  a  smaller  dimension  over  the  coil. 

DESIGN  PROCEDURE  is  carried  out  by: 

(a)   Choosing  a  core  material  and  core  size. 

(6)  Choosing  a  wire  size,  and  determining  how  many  turns  of  this  wire  will  fit  in  the 
core  window.  About  5  per  cent  allowance  should  be  made  for  wires  not  lying  tightly 
together.  Also,  the  total  calculated  build  of  the  coil,  including  spool,  layer  insulation,  and 
outside  wrapper,  should  not  exceed  90  per  cent  of  the  core  window  height. 

(c)  Calculating  the  inductance,  resistance,  core  loss,  heating,  capacity,  and  Q  from  the 
dimensions  and  the  number  of  turns.  Three  or  four  trial  designs  may  be  necessary  before 
the  desired  constants  are  arrived  at. 

Inductance.  Since  sheet  material  cores  are  characterized  by  high  permeability,  prac- 
tically all  the  magnetic  flux  is  confined  within  the  core  structure.  The  inductance  of  a 
sheet-core  inductor,  without  air  gaps,  is  given  by 

x  10-     henry  (1) 

in  which  N  is  the  number  of  turns  on  the  .coil,  A  is  the  cross-section  of  the  core  in  square 
centimeters,  k  is  the  stacking  factor,  I  is  the  mean  length  of  the  magnetic  circuit  in  centi- 
meters, and  juac  is  the  a-c,  or  "incremental,"  permeability  of  the  core  material. 
When  air  gaps  are  present  in  the  core,  the  inductance  is  given  by 

g  x  10-»    henry  (2) 


in  which  ^avg  is  the  average  permeability  of  the  core,  including  air  gaps,    This  average 
permeability  is 

Mavg  —  -,    ,    /_/7^  -  (3) 


in  which  a  is  the  total  effective  length  of  all  air  gaps,  in  centimeters. 

At  any  air  gap  the  magnetic  flux  spreads,  so  that  the  cross-section  of  the  magnetic  field 
is  greater  than  the  cross-section  of  the  core.  It  is  most  convenient  to  treat  this  "fringing" 
as  if  the  length  of  the  air  gap  were  effectively  reduced.  If  m  and  n  are  the  dimensions  of 
the  core  cross-section  at  the  air  gap,  and  lg  is  the  actual  physical  length  of  the  air  gap, 

Effective  length  —  -  -  —  —  —  —  —  X  lg     approximately  (4) 

(m  +  lg)  (n  +  lg) 

Permeability.  Incremental,  or  a-c,  permeability  is  the  kind  of  permeability  of  interest 
in  connection  with  most  inductors.  See  Magnetic  Circuits  and  Transformers,  p.  198.  It 
is  a  variable,  depending  upon: 

(a)  The  material  of  the  core. 

(6)   The  amount  of  d-c  magnetization, 

(c)  The  amount  of  alternating  flux. 

(d)  Wave  form  of  the  a-c  voltage. 

(e)  Previous  magnetization  of  the  core. 
(/)  Temperature. 

Of  these  factors,  only  the  first  three  are  considered  in  practical  design  work,  although  the 
others  are  by  no  means  negligible. 
When  a  core  has  no  air  gaps,  the  d-c  magnetization,  H^c,  is  given  by 

jETdc  =  ~~~l  -    oersteds  (or  gilberts  per  cm.)  (5) 

in  which  I  is  the  direct  current  flowing  through  the  coil,  in  amperes. 


LOW-FREQUENCY,   SHEET-COEE  INDUCTORS 


3-45 


It  is  usually  more  convenient  to  express  the  amount  of  a-c  magnetization  of  the  core  in 
terms  of  flux-density  variation,  which  is  a  function  of  the  a-c  voltage  across  the  coil,  rather 
than  in  terms  of  magnetizing  force,  which  is  a  function  of  the  a-c  current  in  the  coil.  The 
peak  a-c  flux  density,  -Bmax,  is  given  by  eq.  (6),  if  the  a-c  voltage  is  sinusoidal. 


E 


X  10s     gausses 


(6) 


In  this  equation,  E  is  the  rms  voltage  across  the  coil,  /  is  the  frequency,  and  the  other 

symbols  have  the  same  meaning  as      10000 

given  previously  for  eq.  (1).     Equa-          '      r 

tion  (6)  applies  whether  or  not  there 

are  air  gaps  in  the  core.     See  Figs. 

4,  5,  and  6. 

D-c  Magnetization  with  Air  Gaps. 
D-c  magnetization  can  be  reduced  by 
inserting  an  air  gap  in  the  core,  im- 
proving the  a-c  permeability.  See 
Figs.  4,  5,  and  6.  Up  to  a  certain 
point,  this  results  in  an  increase  of 
inductance.  See  eqs.  (2)  and  (3). 
Beyond  that  point,  any  further  in- 


100 


10    20       50    100  500  1000 

A-c  Flux  density,  Bmax,  In  gausses 


FIG.  4.     A-c  Permeability  3.6%  Silicon  Steel  "58"  Grade 


10,000 


crease  in  the  air  gap  causes  a  de- 
crease of  inductance. 

The  amount  of  d-c  magnetization 
in  the  ferromagnetic  core  material,  when  there  is  an  air  gap  in  the  core,  can  be  determined 
by  the  graphical  method  shown  in  Fig.  7.    See  Karapetoff,  The  Magnetic  Circuit.    This 

method  utilizes  the  normal  magneti- 

20,000  i 1    i  i  i  i  mi       i    i  i  i  1 1  in 1  ^r  i  *KJ  1 1 1 1       r      zation  or  B-H  curve  of  the  particular 

core   material,    of  which  several   are 
given  in  Fig.  8. 

The  d-c  magnetization  of  the  core 
material  and  the  a-c  flux  density 
[eq.  (6)]  having  been  determined, 
the  a-c  permeability  is  then  found 
from  curves  such  as  Figs.  4,  5,  or  6. 
This  value  of  permeability  is  used 
in  eq.  (3)  to  compute  the  average 
permeability  of  the  core,  including 
air  gaps,  which  is  then  used  in  eq.  (2) 
to  calculate  the  inductance  of  the 
reactor.  It  is  usually  necessary  to 
try  two  or  three  values  of  air  gap 
to  discover  the  optimum  one.  This 
method  of  determining  the  optimum 
air  gap,  and  the  maximum  inductance,  may  be  called  the  "fundamental  method." 

A  short-cut  method  of  determining  optimum  air  gap  and  maximum  inductance  was 
worked  out  by  C.  R.  Hanna  (De-  2o  000 
sign  of  Reactances  and  Transform- 
ers Which  Carry  Direct  Current, 
Trans,  A.I.E.E.,  February  1929). 
He  showed  that  a  curve  can  be 
drawn,  for  any  particular  core  ma- 
terial, whose  coordinates  are  NI/l 
and  LI2/V,  V  being  the  volume  of 
the  core.  Such  "optimum  design 
curves"  are  shown  for  three  com- 
monly used  materials  in  Figs.  9, 
10,  and  11.  When  using  these 
curves  it  should  be  remembered 
that  they  apply  only  to  a  special 
case,  in  which  maximum  induct- 
ance for  a  given  amount  of  direct 
current  is  the  quality  desired.  If 


200 
100 

10  20   50  100     500  1000       10,000 

A-c  Flux  density,  Bmax,  in  gausses 

FIG.  5.     A-c  Permeability  50-50  Nickel-iron  Alloys 


10,000 


5000 


3000 


2000 


1000 


the  reactor  is  to  be  used  at  several 
different  values  of  direct  current, 


100          200  500          1000 

A-c  Ftux  djenslty,  Bmax,  in  gausses 


2000 


FIG.  6.     A-c  Permeability  Mumetal 


3-46 


RESISTORS,  INDUCTORS,   CAPACITORS 


or  if  other  considerations  govern  the  design,  the  approach  described  previously  should 
be  used. 

Saturation.    If  the  sum  of  the  d-c  flux  density  and  the  peak  a-c  flux  density  approaches 
the  saturation  density  of  the  core  material,  serious  wave-form  distortion  occurs.    This  sum 


° (gausses) 


itization. 


itralght  line 


(oersteds) 


Q.47TNI 


FIG.  7.     Graphical  Method  of  Determining 

D-c  Magnetization  of  Core,  when  there  is  an 

airgap 


12345 
H  magnetizing  force  in  oersteds 

FIG.  8.     D-c  Magnetization  Curves 


should  not  exceed  12,000  gausses  for  silicon  steel  or  for  50-50  nickel  alloys.    The  d-c  flux 
density  may  be  found  by  the  graphical  method  of  Fig.  7,  and  the  peak  a-c  flux  density, 


0.0026 
0.0024 
0.0022 
0.0020 
0.0018 
0.0016 
LI20.0014 

v 

0.0012 
0.0010 
0.0008 
0.0006 
0.0004 
0.0002 
0 

0.0  4^ 

< 

i     /T 

// 

^ 

o.oc 

m 

003 

% 

'i 

0.0 

K" 

0.0 

025 

^ 

A  ?-00 

20V 

7f.cc 

30 

*  0.0015,/ 

D02! 

. 

/ 

^ 

•«  r\ 

320 

D.OO 

10X 

'  ^ 

^ 

/ 

/ 

/  0 

001 

5 

UUOj 

{S 

"o.c 

010 

But 

:  joir 

^f*C 

.00 

15 

^ 

^ 

Butt  joint. 

Dim 

ensii 

>ns  i 

i  ce 

itim< 

ters 

0  5  10  15  20  25  30  35 

NI/Z 
FIG.  9.     Optimum  Design  Curves  for  Inductance  with  D.C.  3.6%  Silicon 

-Bmax,  from  eq.  (6) .  If  the  total  flux  density  is  excessive,  and  the  d-c  flux  density  is  the 
larger  part,  the  air  gap  should  be  increased.  If  the  a-c  flux  density  is  the  larger  part,  the 
number  of  turns  on  the  coil  should  be  increased. 


LOW-FREQUENCY,   SHEET-CORE   INDUCTORS 


3-47 


0.0030 
0.0028 
0.0026 
0.0024 
0.0022 
0.0020 

0.0018 

II! 
V    0.0016 

0.0014 
0.0012 
0.0010 
0.0008 
0.0006 
0.0004 
0,0002 
0 

H 

04, 

04 

003 

0 

7, 

i 

R 

nax 
gau 

=100 

sses 
003 

A 

/t 

X).C 

035 

V 

-«, 

^ 

O.OC 

25  , 

//\™\ 

33 

,°* 

y^ 

Trax 
gau£ 

=10 
ses 

3 

<, 

^    0 

002 

'vo 

002 

^ 

/< 

y 

n  or 

.00 

f5X 

r 

.00: 

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001 

5 

^ 

3.00 

35 

/" 

o.oc 

r 

$ 

1.00 

^ 

But 

join 

iX 

^ 

/[But 

:|o! 

ilt 
1 

Dim 

ms 

ns 

i  cer 

tlm 

ters 

3              5             10            15             20            25            30            35 

NI/Z 
FIG.  10.     Optimum  Design  Curves  for  Inductance  with  D.C.  50-50  Nickel-iron  Alloys 


0.0038 
0.0036 
0.0034 
0.0032 
0.0030 
0,0028 
0,0026 
0.0024 
0.0022 
V   0.0020 
0.0018 
0.0016 
0.0014 
0.0012 
0.0010 
0.0008 
0.0006 
0-0004 
0.0002 

0 

C 

xoossyj      | 

\// 

X 

~ 
303 

^ 

0-0 

330 

/ 

•X 

^ 

/<£ 

f  o.oos 

5/ 

^ 

^0 

003 

3 

/ 

/ 

C? 

/ 

/ 

0.0 

D20 

/ 

/o. 

302, 

) 

/ 

/ 

// 

0 

001 

y 

/o 

.002 

0 

/ 

y 

0.0 

310 

// 

0.0 

015 

/, 

/ 

-O.C 

nop 

/ 

r  ° 

.00: 

0 

A 

K 

o.oc 

tt  Jo 

^ 

^BL 

poo 

int 

Dim* 

nslo 

ns  ir 

cen 

time 

:ers 

)               5              10             15         .  20            25             30            3E 
NI/Z 

FIG.  11.     Optimum  Design  Curves  for  Inductance  with  D.C.  Hipersil 


3-48 


RESISTORS,   INDUCTORS,   CAPACITORS 


RECTIFIER-FILTER  REACTORS  AND  AUDIOFREQUENCY  PLATE  REACTORS 

are  examples  of  inductors  for  which  maximum  inductance  at  a  given  value  of  direct  current 
is  the  most  important  quality  desired.  The  "optimum  design  curves"  are  ideally  suited 
for  such  designs. 

THE  SWINGING  CHOKE  is  a  reactor  which  must  have  a  specified  inductance  at  some 
large  value  of  direct  current,  but  which  must  increase  rapidly  in  inductance  as  the  direct 
current  is  decreased.  An  air  gap  is  used  which  is  optimum,  not  for  the  largest  direct  cur- 
rent nor  for  the  smallest  direct  current,  but  rather  for  some  intermediate  value.  An  ap- 
proximate design  can  be  arrived  at  by  using  the  "optimum  design  curves"  for  an  interme- 
diate value  of  direct  current.  Then  the  inductance  should  be  calculated  for  the  maximum 
and  minimum  currents  by  the  fundamental  method.  In  some 
cases,  part  of  the  core  stack  may  have  a  large  air  gap  to 
provide  a  good  inductance  at  the  largest  current,  while  the 
remainder  of  the  core  may  have  a  small  air  gap  to  give  high 
inductance  at  the  minimum  current. 

SATURABLE  REACTORS  are  inductors  the  inductance  of 
which  is  varied  by  means  of  a  variable  direct  current.  Inter- 
leaved laminations  are  used  to  get  the  greatest  effect  of  the 
d-c  magnetization.  It  is  necessary  to  isolate  the  d-c  circuit 
from  the  a-c  circuit  by  means  of  separate  windings.  It  is  also 
necessary  to  prevent  transformer  coupling  between  the  a-c 
and  d-c  windings.  Figure  12  illustrates  the  construction  of 
a  saturable  reactor.  The  a-c  windings  should  always  be 
See  Holubow,  "D-C  Saturable  Reactors  for  Control  Purposes," 


A-c 

n 

__7 

circuit 

LJ 

—  ' 

\ 

Interleaved  core 

D-c  circuit 

FIG.  12.     Construction  of 
Saturable  Reactor 


connected  in  parallel. 

Electronic  Industries,  March  1945. 

COIL  RESISTANCE.  The  mean  length  of  turn  is  calculated  from  the  coil  geometry, 
converted  into  feet,  and  multiplied  by  the  number  of  turns.  Reference  to  a  wire  table 
(Section  2,  article  3)  will  give  the  d-c  resistance  of  the  coil,  in  ohms.  Skin  effect  can  be 
neglected,  except  for  very  large  wire,  above  1  kc.  Allowance  for  temperature  must  be 
made.  The  resistance  of  a  coil  increases  about  0.4  per  cent  per  degree  centigrade,  above 
20  deg  cent.  Thus,  if  an  inductor  operates 
at  a  coil  temperature  of  70  deg  cent,  its 
resistance  will  be  20  per  cent  higher  than 
calculated  for  20  deg  cent. 

CORE-LOSS  curves  are  shown  in  Sec- 
tion 6,  article  14.  The  core  loss  is  deter- 
mined in  watts,  which  can  then  be  added 
to  copper  loss  watts  to  find  the  total  heat 
dissipated  in  the  reactor,  and  the  tempera- 
ture rise. 

When  the  effect  of  core  loss  on  the  Q 
of  the  inductor,  rather  than  heating,  is  the 
matter  of  interest,  core  loss  can  be  repre- 
sented by  a  resistor  in  parallel  with  the 
coil,  of  a  value  Ri  ohms. 


W 


(7) 


1 

-o 

e; 

a 

a. 

V) 

0.1 

—  J 

"7 

t- 

// 

^ 

i 

// 

^ 

/ 

sff 

/ 

/ 

J 

^ 

|v 

/ 

r" 

/ 
^ 

5 

i 

i^^zz 

>v     / 

*s 

I 

N' 

sr  /ff 

^•j 

*!/ 
N 

/j 

/ 

r  / 

xf< 
/"V 

f 

0.01 
5 

/f/~ 

7~7 

/    / 

/ 

/ 

/    // 

/x 

^ 

*/ 

0 

100                              1000                           10,000 

In  this  formula,  E  is  the  rms  voltage 
across  the  coil,  and  W  is  the  watts  core 
loss.  The  core  loss  of  most  sheet-core 
materials  varies  very  nearly  as  the  square 
of  the  applied  voltage,  so  that  the  ratio 
E2/W  is  very  nearly  constant  over  a  con- 
siderable range  of  voltage.  Figure  13 
gives  core-loss  curves  of  several  materials 
at  2000  gausses.  If  P  is  the  watts  per 
pound,  at  2000  gausses,  and  M  is  the  weight  of  the  core,  in  pounds,  the  core-loss  resistance  is 


Cycles  per  aecond 
FIG.  13    Core  Loss  at  2000  Gausses  CBmax) 


3~10    ohm 


(8) 


OPTIMUM  Q.  Q  is  often  defined  as  the  ratio  of  reactance  to  resistance.  This  is  true 
when  the  only  resistance  present  is  in  series  with  the  coil,  as  is  the  case  with  the  copper 
resistance  of  the  winding.  When  there  are  both  series  and  shunt  resistances,  as  with  a 


LOW-FREQUENCY,   SHEET-CORE  INDUCTORS  3-49 


ferrous-cored  inductor,  Q  is  more  clearly  defined  as  the  ratio  of  reactive  volt-amperes  to 
resistive  volt-amperes. 

For  a  particular  core  and  coil,  at  a  particular  frequency,  the  shunt  resistance  of  the 
core  and  the  series  resistance  of  the  coil  are  practically  fixed  quantities.  The  reactance, 
however,  can  be  varied  by  changing  the  air  gap.  There  is  a  value  of  reactance  which  will 
produce  the  maximum  Q.  This  problem  is  illustrated  by  the  circuit  shown  in  Fig.  14.  In 
this  circuit,  Rc  is  the  resistance  of  the  coil,  Ri  is  the  resist- 
ance of  the  core,  and  X  is  the  reactance  of  the  inductor. 
Maximum  Q  is  attained  when  CoU  resistance  j 

•^-^    or    X  =  VRJ2i  (9)  Ri| 

•"'C  -^  Core-loss  1 

resistance^ 

The  air  gap  is  adjusted  so  that 


X 

Col! 
reactance 


27T/ 


for  maximum  Q 


(10)     FIG.    14.     Equivalent   Circuit, 
Ferrous-cored  Inductor 


Equations  (2)  and  (3)  are  used  to  compute  the  correct  air  gap.    When  the  air  gap  and 
inductance  are  adjusted  to  optimum, 

V 


The  maximum  Q  that  can  be  obtained  in  a  low-frequency,  sheet-core  inductor  is  prac- 
tically independent  of  the  inductance,  the  number  of  turns,  the  voltage,  or  the  flux  density. 
That  is,  a  10-henry  reactor  can  be  made  with  practically  the  same  Q  as  a  1-mh.  reactor, 
by  proper  design.  Maximum  Q  that  can  be  realized  depends  only  on  the  core  material, 
the  size  of  the  core,  and  the  frequency.  See  Fig.  15.  For  other  sizes  of  core,  the  Q  will 


10 


200 


5000     10,000 


500        1000       2000 
Cycles  per  second 

FIG.  15.     Q  vs  Frequency,  Various  Core  Materials 

vary  as  the  linear  dimension  if  the  proportions  are  the  same,  or  as  the  cube  root  of  the  core 
volume  if  the  proportions  are  different.  Thus,  a  core  having  twice  the  linear  dimensions 
of  the  one  shown  will  have  8  times  the  volume  but  will  give  only  twice  the  Q.  For  other 
core  materials,  the  Q  that  can  be  realized  will  vary  inversely  as  the  square  root  of  the 
relative  core  loss.  Thus,  a  material  having  one-fourth  the  core  loss  will  give  twice  the  Q. 

Air  Gaps  for  Q  should  be  located  inside  the  coil,  and  preferably  at  the  center  of  the  coil 
length.  There  should  be  no  air  gaps  outside  of  the  coil.  Fringing  flux  creates  a  magnetic 
field  external  to  the  reactor.  When  the  reactor  is  placed  in  a  metal  can,  or  near  a  metal 
chassis,  eddy  currents  are  set  up,  which  may  reduce  the  Q  of  the  reactor  by  a  factor  of 
2  or  3  to  1. 

Optimum  Permeability  for  Q.  For  a  particular  core  and  coil,  at  a  particular  frequency, 
there  is  an  optimum  value  of  inductance  which  will  give  the  highest  Q.  See  eq.  (10) .  This 
value  of  inductance  requires  a  certain  average  permeability  of  the  core.  See  eq.  (2). 


3-50 


RESISTORS,  INDUCTORS,   CAPACITORS 


1000 


Practically  all  sheet-core  materials  have  too  high  a  value  of  a-c  permeability  to  give  the 
optimum  inductance,  so  that  it  is  necessary  to  reduce  the  overall  permeability  of  the  core 
by  inserting  an  air  gap.  The  use  of  an  air  gap  to  produce  maximum  Q  in  an  inductor 

should  not  be  confused  with  the  use  of 
an  air  gap  to  reduce  d-c  magnetization. 
The  two  purposes  are  different  and  dis- 
tinct. 

The  optimum  overall  permeability, 
Pavg,  required  to  produce  maximum  Q  is 
a  function  of  the  core  material,  the  core 
size,  and  the  frequency.  See  Fig.  16. 
For  other  sizes  of  core,  optimum  perme- 
ability will  vary  inversely  as  the  linear 
dimensions  if  the  proportions  are  the 
same,  or  inversely  as  the  cube  root  of  the 
core  volume  if  the  proportions  are  dif- 
ferent. For  other  core  materials,  the 
optimum  permeability  will  vary  in- 
versely as  the  square  root  of  the  relative 
10,000  core  loss. 

A  striking  fact  brought  out  by  Fig.  16 
is  that  the  optimum  overall  permeability 
becomes  less  than  50  at  frequencies 


50       100 


FIG.  16. 


1000 
Cydes  per  second 

Permeability  for  Maximum  Q 


above  4000  cycles.  The  high  a-c  permeability  characteristic  of  sheet-core  materials  is  of  no 
advantage  in  this  region.  Permeability  of  the  order  of  50  can  be  realized  with  powdered- 
iron  cores.  This  opens  up  the  possibility  of  using  powder  cores  at  such  frequencies. 


16.  HIGH-FREQUENCY,  POWDERED-CORE  INDUCTORS 

Powdered  cores  differ  from  sheet  cores  in  two  important  characteristics.  They  have, 
in  general,  lower  core  loss,  particularly  eddy-current  loss,  and  they  have  much  lower 
permeability.  Both  these  characteristics  are  due  to  the  subdivision  of  the  core  into  minute 
particles,  and  both  are  desirable  at  high  frequencies  if  the  Q  of  the  inductor  is  a  major 
consideration.  See  above,  "Optimum  Permeability,"  and  Fig.  16. 

Material  used  for  powdered  cores  are  magnetite,  a  natural  iron  oxide,  electrolytic  iron, 
hydrogen-reduced  iron,  carbonyl  iron,  powdered  Permalloy,  and  powdered  molybdenum 
Permalloy.  For  descriptions  of  these  materials  and  their  method  of  manufacture,  see 
Section  2,  "Magnetic  Materials";  H.  G,  Shea,  "Magnetic  Powders,"  Electronic  Industries, 
August  1945;  V.  E.  Legg  and  F.  J. 
Given,  "Powdered  Molybdenum 
Permalloy,"  B.S.T.J.,  July  1940. 

Particle  sizes  range  from  about  50 
microns  diameter  (0.002  in.)  for  audio- 
frequency cores  to  about  3  microns  for 
cores  useful  at  100  me.  The  particles 
making  up  a  core  are  not  all  the  same 

size.    This  is  an  advantage,  as  it  gives     (a\  R^  core  tc\  SIu 

a  better  packing  of  the  magnetic  mate- 
rial,  the  smaller  particles  filling  in  the 
spaces  between  the  larger  ones.  Since 
eddy-current  loss  is  proportional  to  the 
square  of  the  particle  diameter,  the 
root-mean-square  diameter  of  the  par- 
ticles making  up  a  core  is  the  particle 
dimension  of  greatest  significance. 
Weight-average  diameter  is  often 
given.  This  is  a  figure  a  few  per  cent 
higher  than  the  root-mean-square  di- 
ameter. 


(d)  Core  arrd  shell  (e)  Oore  and  half  shaQs 

FIG.  17.     Basic  Types  of  Powdered-iron  Cores 


CORE  CONSTRUCTION.  Particles  are  coated  with  an  insulating  material,  or  treated 
in  some  manner  to  give  them  high  surface  resistivity,  and  mixed  with  a  plastic  binder. 
The  mixture  is  compressed  in  molds  at  pressures  of  50  to  100  tons  per  square  inch,  to  form 
the  desired  core  shape.  The  resulting  core  is  a  solid  block  of  material  resembling  iron  in 
appearance  and  somewhat  lighter  than  iron  in  weight.  Figure  17  illustrates  a  few  basic 


HIGH-FREQUENCY,   POWDERED-CORE  INDUCTORS      3-51 


styles.  As  better  audiofrequency  core  materials  are  developed,  cores  of  the  shapes  shown 
in  Fig.  1  are  also  being  offered. 

COIL  CONSTRUCTION,  in  general,  follows  the  practice  used  for  air-core  coils  of  the 
same  frequency  range.  Since  powdered-iron  cores  are  employed  at  the  higher  audio- 
frequencies and  above,  skin  effect,  coil  capacitance,  and  coil  dielectric  loss  are  controlling 
factors  in  coil  design,  as  with  air-core  coils.  See  Section  2,  "Inductors  with  Air  Cores." 
The  presence  of  a  core  introduces  two  modifications  of  coil  design.  A  somewhat  longer 
coil,  of  less  build-up,  is  desirable,  to  take  better  advantage  of  the  core.  Dielectric  loss  and 
dielectric  constant  of  the  winding  form  are  of  more  importance  when  there  is  a  core, 
especially  if  the  core  is  grounded. 

PERMEABILITY.  "Intrinsic  permeability"  is  the  permeability  of  the  magnetic 
particles.  "Composite  permeability,"  "ring-core  permeability,"  or  just  "permeability'' 
are  terms  used  to  describe  the  permeability  of  a  core,  made  in  the  form  of  a  ring  sample, 
and  wound  with  a  toroidal  coil.  Such  a  sample  has  uniform  flux  density  throughout  its 
length  and  practically  throughout  its  cross-section,  and  no  end  effects.  Consequently, 
the  ring-core  permeability  may  be  called  the  true  permeability  of  the  core  material.  The 
relation  between  ring-core  permeability,  ju,  and  intrinsic  permeability  juz-,  is  given  by 

M  «  (Mf)p  (12) 

in  which  p  is  the  "packing  factor,"  or  fraction  of  the  core  volume  occupied  by  magnetic 
material.  See  Legg  and  Given,  also  analysis  by  H.  Beller  and  G.  O.  Altmann,  "Radio- 
Frequency  Cores  of  High  Permeability,"  Electronic  Industries,  November  1945.  Table  3 
gives  values  of  ring-core  permeability  for  a  number  of  core  materials. 

The  term  "effective  permeability"  is  used  to  describe  the  permeability  of  a  cylindrical 
or  slug-type  core  [Fig.  17(&)].  Such  permeability  is  much  lower  than  ring-core  permeability 
of  the  same  core  material,  be- 
cause of  demagnetizing  effects 
at  the  ends  of  the  cylinder. 
This  matter  has  been  investi- 
gated by  R.  M.  Bozorth  and 
D.  M.  Chapin,  "Demagnetiz- 
ing Factors  of  Rods,"  J.  Ap- 
plied Phys.,  May  1942,  and 
by  W.  J.  Polydoroff  and  A.  J. 
Klapperich,  "Permeability  of 
High  Frequency  Iron  Cores," 
Radio,  November  1945.  Ta- 
ble 2  shows  effective  perme- 
ability versus  ring-core  per- 
meability for  various  ratios  of 
core  length  to  core  diameter. 

INDUCTANCE. 


Table  2.     Effective  Permeability  of  Cylindrical  Cores 


Ring-core 
Permeability 

Ratio  Length/Diameter 

1 

2 

4 

6 

8 

Effective  Permeability 

5 
10 
15 
20 
25 
30 

2.20 
2.98 
3.47 
3.76 
3.94 
4.06 

3.40 
4.57 
5.65 
6.3 
6.6 
6.75 

4.1 
6.8 
8.7 
9.9 
10.6 
11.0 

4.4 
8.2 
10.8 
12.4 
13.4 
14.0 

4,8 
8.8 
11.7 
13.7 
14.8 
15.6 

The  coil  is  the  same  length  as  the  core,  in  these  data. 
The  inductance  of  a  toroidal  coil  on  a  ring  core  is  given  by 


4.  AT2 

=j 

a 


. 
~=j-[A'  +  A.(p  -  1)]  X  10'fl     henry 


(13) 


in  which  N  is  the  number  of  turns  on  the  coil;  d  is  the  mean  diameter  of  the  core,  in  centi- 
meters; A'  is  the  mean  cross-section  of  the  coil,  at  right  angles  to  the  flux  path,  in  square 
centimeters;  A  is  the  cross-section  of  the  core,  in  square  centimeters;  and  AC  is  the  ring-core 
permeability  of  the  core. 

The  inductance  of  a  coil  on  a  cylindrical,  or  slug-type  core,  if  the  coil  is  the  same  length 
as  the  core,  is 


In  this  equation,  Z/o  is  the  inductance  of  the  coil  without  iron,  6t-  is  the  radius  of  the  iron 
core,  60  is  the  mean  radius  of  the  coil,  and  p*  is  the  effective  permeability,  as  given  in 
Table  2.  If  the  core  is  longer  than  the  coil,  the  effective  permeability  is  increased  to  a  value 


At.' 


(15) 


k  being  the  length  of  the  core,  and  Z0  that  of  the  coil.    This  higher  value  should  be  sub- 
stituted for  ne  in  eq.  (14) . 

Formulas  are  not  available  which  apply  to  the  various  shell-type  cores.    The  variety 
of  shapes,  non-uniform  cross-section  of  the  core,  and  square  corners  in  the  magnetic  path 


3-52 


RESISTORS,   INDUCTORS,    CAPACITORS 


offered  by  the  core  make  any  accurate  analysis  very  difficult.  The  inductance  of  a  coil, 
having  a  central  core  and  a  shell,  will  be  greater  than  its  inductance  with  the  central  core 
alone.  On  the  other  hand,  its  inductance  will  be  less  than  that  of  a  toroidal  coil,  of  the 
same  number  of  turns,  on  a  ring  core  of  the  same  average  cross-section  and  the  same  mean 
length  of  magnetic  path.  See  "Measurement  of  Iron  Cores  at  Radio  Frequencies,"  D.  E. 
Foster  and  A.  E.  Newlon,  Proc.  I.R.E.,  May  1941,  and  above  reference  by  Polydoroff  and 
Klapperieh. 

CORE  LOSS  AND  Q.  Core  loss  is  of  interest  in  high-frequency  inductors  primarily 
because  of  its  effect  upon  Q,  or  quality  factor.  The  addition  of  the  core  increases  not  only 
inductance  but  also  resistance.  A  toroidal  coil  on  a  ring  core  has  been  analyzed  by  V.  E. 
Legg,  "Magnetic  Measurements  at  Low  Flux  Densities,"  B.S.T.J.,  January  1936.  His 
formula  for  the  increase  of  resistance  due  to  core  loss  is 


R  =  [(aBm  +  c)/  + 


ohms 


(16) 


In  this,  R  is  the  resistance  added  to  a  coil  by  the  core;  a,  c,  and  e  are  the  hysteresis,  residual, 
and  eddy-current  loss  coefficients;  /is  the  frequency,  in  cycles  per  second;  Bm  is  the  peak 
a-c  flux  density,  in  gausses;  /x  is  the  ring-core  permeability;  and  L  is  the  inductance  of  the 
coil  with  the  core,  in  henries.  Coefficients  are  given  in  Table  3  for  a  number  of  core 
materials. 

The  resistance  added  to  a  coil  by  a  cylindrical  core  has  been  analyzed  by  Foster  and 
Newlon,  reference  above,  who  give  a  formula 


••L, 


bo2  V4b02  +  tf 


ohms 


(17) 


Symbols  have  the  same  meaning  as  for  eqs.  (14)  and  (15),  p  being  the  loss  factor  of  the 
core  material.    The  loss  factor  must  be  determined  by  measurements  upon  a  sample  of 

the  particular  core  mate- 
Table  3.     Ring  Permeability  and  Core-loss  Coefficients         rial.     Published  data  are 

not  available  on  the  loss 
factor  of  cylindrical  cores. 
In  Table  3,  the  figures 
for  the  permalloys  are 
taken  from  Legg  and 
Given.  Figures  for  the 
other  materials  are  Gen- 
eral Aniline  and  Film  Cor- 
poration data,  published 
in  the  articles  by  H.  G. 
Shea,  and  by  Beller  and 
Altmann,  previously  re- 
ferred to. 

PERMEABILITY 
TUNING,  or  "variable  re- 
luctance tuning,"  is  the 
system  of  adjusting  a  cir- 
cuit to  resonance,  at  a  de- 
sired frequency,  by  mov- 
ing an  iron  core  in  or  out 
of  the  coil.  A  simple  ar- 
rangement for  doing  this 
is  to  mold  a  screw  in  one 
end  of  a  cylindrical  core  as  illustrated  in  Fig.  17 (c).  The  screw  passes  through  a  tapped 
hole  in  the  coil  housing  and  is  slotted  at  the  end  for  a  screwdriver.  Rotation  of  the  screw 
moves  the  core  axially  into  or  out  of  the  coil,  varying  its  inductance  and  the  resonant 
frequency  of  the  circuit.  Such  a  system  is  ideal  for  a  circuit  tuned  to  some  fixed  frequency, 
such  as  an  intermediate-frequency  transformer  circuit.  See  "Ferro-inductors  and  Perme- 
ability Tuning,"  W.  J.  Polydoroff,  Proc.  I.R.E.,  May  1933. 

Incremental  permeability  tuning  is  a  system  of  adjusting  the  resonant  frequency  of  a 
circuit  by  varying  the  permeability  of  the  iron  core  without  any  mechanical  motion.  The 
permeability  is  varied  by  means  of  d-c  magnetization  on  the  same  principle  as  the  saturable 
reactor.  Increase  of  d-c  magnetization  causes  a  decrease  of  inductance  and  an  increase 
of  frequency.  By  proper  design,  the  increase  of  frequency,  from  some  minimum  value, 
may  be  made  proportional  to  the  direct  current.  "Incremental  Permeability  Tuning," 
W.  J.  Polydoroff,  Radio,  October  1944. 


Material 

Perme- 
ability 

Hysteresis 
t*a  X  103 

Residual 
MC  X  103 

Eddy 
Current 

IJ.6  X    106 

2-8  1  Molyb.  Perm  
2-81  Molyb.  Perm  
2-81  Molyb.  Perm  
8  1  Permalloy 

125 
26 
14 
75 

0.20 
0.18 
0.16 
0  41 

3.8 
2.5 
2,0 
2  8 

2.4 
0.2 
0.1 
3  8 

81  Permalloy 

26 

0.30 

2.8 

0  7 

Carbonyl  "55".  . 

55 

0.86 

18.0 

0  073 

Carbonyl  L     

39 

1.45 

27.0 

0.  10 

Carbonyl  L 

24  8 

3  1 

0   13 

Carbonyl  C 

16  7 

1   i 

0   14 

Carbonyl  E     

10.4 

0.3 

0.  11 

Carbonyl  TH  

9.6 

0.3 

0.  10 

Carbonyl  SF 

8   1 

0  3 

0   10 

Electrolytic 

23  4 

2.4 

0  33 

Hydrogen-reduced.  .  . 
Hydrogen-reduced.  .  .  . 
Hydrogen-reduced.  .  .  . 
Hydrogen-reduced.  .  .  . 
Magnetite       

42 
18.4 
16.9 
12.5 
7  9 

1.04 
2.6 
1.0 
3.1 
9.  1 

23.0 

0.17 
0.12 
0.12 
0.11 
11.5 

Magnetite  

5.7 

6.8 

0.21 

Magnetite  

3.1 

0.3 

0.085 

CLASSIFICATION  OF  CAPACITOKS 


3-53 


CAPACITORS 

By  James  I.  Cornell 

A  capacitor  or  condenser  is  an  electrical  device  used  primarily  because  it  possesses  the 
property  of  capacitance.  Though  capacitor  is  the  preferred  engineering  terminology,  the 
term  condenser  is  still  widely  used. 

Electrostatic  capacitance  is  denned  as  the  ratio  of  the  electrical  charge  Q  stored  in  the 
capacitor  by  virtue  of  the  applied  voltage  (J?dc) .  That  is,  when  a  d-c  voltage  is  impressed 
on  two  conductors  insulated  from  each  other,  the  voltage  causes  an  electrical  charge  to 
flow  into  the  system.  One  conductor  assumes  a  positive  charge  and  the  other  an  equal 
negative  charge,  depending  upon  the  polarity  of  the  impressed  voltage.  The  charge  on 
the  conductors  produces  electrostatic  stresses  in  the  region  between  them.  The  work 
done  in  charging  the  capacitor  appears  as  stored  potential  energy.  This  energy  is  released 
when  we  remove  the  impressed  voltage  and  short-circuit  the  capacitor  electrodes.  The 
capacitance  is  a  measure  of  the  charge  or  stored  potential  energy  for  a  given  voltage.  In 
terms  of  physical  units,  a  capacitor  having  a  capacitance  of  1  farad  will  store  1  coulomb  of 
charge  or  1  watt-second  of  energy  for  1  volt  of  applied  direct  voltage. 


17.  CLASSIFICATION  OF  CAPACITORS 

Capacitors  may  be  classified  according  to  form,  dielectric  medium,  and  electrode  structure. 
Generally,  the  classification  of  capacitors  according  to  form  is  determined  by  whether 
the  capacitance  is  variable  or  fixed. 

Variable  capacitors  cover  two  essential  types,  namely: 

1.  Variable  capacitors  provide  for  continuous  control  of  the  capacitance  and  are  used 
for  varying  the  resonance  frequency  of  a 

tuned  circuit  with  which  it  is  associated. 
The  dielectric  employed  may  be  air,  com- 
pressed gas,  or  liquid  types.  See  Fig.  1. 

2.  Adjustable  capacitors  provide  for  lim- 
ited control  of  the  capacitance,  and  these 
capacitors  are  usually  found  in  frequency- 
determining  circuits  for  alignment  pur- 
poses   such    as    intermediate-frequency 
transformers,  etc.    The  dielectric  medium 
used  in  adjustable-type  capacitors  may 
be  air,  mica,  or  some  form  of  ceramic. 

Fixed  capacitors  employ  a  wide  range 
of  dielectric  materials,  and,  because  of  the 
different  dielectric  media  used,  the  form 
and  usage  of  these  capacitors  must  be 
evaluated  according  to  their  dielectric  in 
terms  of  circuit  requirements.  The  wide 
variation  in  the  kinds  of  dielectric  mate- 
rials available  for  capacitors  permits  con- 
struction to  meet  practically  all  kinds  of 
circuit  requirements. 

FIXED  CAPACITORS  CLASSIFIED  ACCORDING  TO  DIELECTRIC  MEDIUM. 
Gas  Dielectrics.  Vacuum  and  compressed-gas-filled  capacitors  are  designed  for  elec- 
tronic power  circuits  where  high  voltage,  high  current,  and  high  frequency  requirements 
are  encountered  and  where  other  types  of  dielectrics  are  inadequate  because  of  excessive 
losses  and  resultant  overheating. 

Impregnated  Paper.  The  most  common  form  of  solid-dielectric  capacitors  employed 
in  electronic  and  communication  circuits  are  impregnated-paper  types.  High-purity  kraft 
paper  may  be  impregnated  with  micro  crystalline  hydrocarbon  waxes,  chlorinated  waxes, 
vegetable  oil,  mineral  oil,  Askarels  or  synthetic  chlorinated  oils,  and  plastics.  Impreg- 
nated-paper capacitors  are  usually  made  up  in  a  multiple-layer  foil  and  paper  structure 
of  rolled  construction  and  are  impregnated  after  winding. 

Plastic  Film.  Special  design  considerations  such  as  low  dielectric  absorption  or  high 
operating  temperatures  have  resulted  in  the  development  of  plastic-film  dielectrics,  using 
such  substances  as  polystyrene,  acetate,  or  butyrate.  These  follow  the  general  construc- 
tion of  the  impregnated-paper  types  except  that  the  plastic  dielectrics  are  in  final  form  and 


FIG.  1.     Typical  Variable  Capacitors 


3-54  RESISTORS,   INDUCTORS,   CAPACITORS 

are  Impregnated  only  for  the  purpose  of  removing  surface  moisture  and  eliminating  voids 
in  the  winding. 

The  principal  virtue  of  polystyrene  lies  in  the  very  low  absorption  characteristic  and  its 
low  losses  at  radio  frequencies.  Its  use  is  limited  to  an  operating  temperature  of  85  deg 
cent.  Other  synthetic  plastic  films  such  as  acetates  and  butyrates  have  been  found  useful 
for  applications  requiring  operation  at  ambient  temperatures  exceeding  85  deg  cent,  and 
they  have  been  operated  experimentally  at  150  deg  cent.  They  usually  have  high  r-f  power 
factor,  and  their  use  is  restricted  to  d-c  or  low  voltage  a-c  circuits  where  high  ambient 
temperature  is  the  principal  consideration. 

Inorganic  Dielectrics.  Common  inorganic  dielectric  materials  are  solid  dielectrics  con- 
sisting of  mica,  glass,  and  ceramic  types;  also  liquid  dielectrics  such  as  silicone  oils.  The 
solid  dielectrics  are  usually  in  sheet  or  plate  form  which  require  a  laminated  stacked  con- 
struction. The  electrodes  may  take  the  form  of  foil,  or  a  silver  coating  deposited  directly 
on  the  surface  of  the  dielectric  before  the  stacking  operation. 

Ceramic  dielectrics  are  supplied  in  a  variety  of  shapes  other  than  laminal,  and  one  of 
the  most  popular  is  a  hollow  cylinder  with  the  electrodes  placed  on  the  inner  and  outer 
surfaces  in  the  form  of  a  silver  coating.  They  form  a  capacitor  whose  capacitance  is  a 
function  of  the  dielectric  constant  of  the  ceramic.  Capacitance  values  are  usually  low  or 
less  than  2000  n/mi.  The  dielectric  constant  and  temperature  coefficient  of  the  ceramic 
body  can  be  varied  widely  to  give  capacitors  with  negative,  positive,  or  zero  temperature 
coefficients  of  capacitance. 

This  form  of  capacitor  is  receiving  more  consideration  in  r-f  circuits  where  a  negative 
temperature  coefficient  is  used  to  compensate  for  the  positive  temperature  coefficients  of 
other  circuit  elements  with  which  they  are  associated. 

The  electrode  structure  used  with  a  silicone  oil  dielectric  is  of  the  rigid  grid  type  similar 
to  that  found  in  variable  capacitors.  Silicone  oils  will  withstand  high  ambient  tempera- 
tures and  are  characterized  by  low  r-f  power  factor. 

Oxide  Film  Dielectrics.  Electrolytic  capacitors  owe  their  unusual  characteristics  to 
the  oxide  film  dielectric  layer  which  is  produced  electrochemically  on  aluminum,  tantalum, 
and  certain  other  metals.  Details  of  this  form  of  dielectric  will  be  found  under  the  subject 
"Electrolytic  Capacitors." 

CLASSIFICATION  OF  FIXED  CAPACITORS  ACCORDING  TO  PLATE  STRUC- 
TURE. Rigid  multiple  parallel  plate  structure  used  in  connection  with  gas  or  liquid  di- 
electrics. Under  this  classification  may  be  found  the  silicone  and  compressed  gas  capacitors 
for  high-power,  high-frequency  radio  transmitting  and  dielectric  heating  circuits;  also  wet 
electrolytic  capacitors  for  use  in  low-voltage  receiver  power-supply  filters. 

Interleaved  foil  or  stack  construction  used  in  connection  with  mica,  glass,  ceramic,  and 
plastic  film  solid  dielectrics. 

Helical  or  Rolled  Plate  Construction.  The  most  common  form  of  plate  structure  for 
solid  dielectrics  such  as  impregnated  paper,  plastic  film,  and  electrolytic  types  is  the  helical 
or  rolled  plate  construction.  In  this  construction  the  electrodes  are  of  very  thin  foil 
separated  by  a  single-  or  multiple-ply  dielectric  layer  and  wound  into  a  cylindrical  roll. 

Metallized  plate  construction  where  the  electrodes  are  deposited  on  the  surface  of  the 
dielectric.  This  construction  is  used  with  mica,  ceramic,  glass,  plastic,  and  impregnated- 
paper  dielectrics. 

CAPACITOR  CHARACTERISTICS.  Variable  Nature  of  Capacitance.  Three  principal 
characteristics  of  any  dielectric  medium  influence  the  physical  form  of  capacitors: 

1.  Dielectric  constant  or  specific  inductive  capacity,  denoted  by  K. 

2.  Dielectric  strength. 

3.  Dielectric  loss  or  power  factor. 

The  dielectric  constant  is  a  numerical  quantity,  expressed  as  the  ratio  of  the  capacitance 
of  the  structure  with  a  dielectric  other  than  air  to  the  capacitance  with  air  as  the  dielectric. 
In  other  words,  the  higher  the  specific  inductive  capacity  or  dielectric  constant  (K) ,  the 
smaller  the  size  for  a  given  capacitance.  Note:  Refer  to  Section  2  for  the  dielectric  constant 
of  various  materials. 

Dielectric  strength  may  be  defined  as  the  property  of  the  dielectric  by  which  it  with- 
stands breakdown  when  a  voltage  is  applied  to  it.  The  dielectric  strength  is  expressed  in 
volts  per  mil  of  dielectric  thickness.  The  shape  of  the  electrodes  by  which  the  voltage  is 
impressed  on  the  dielectric  and  the  duration  of  the  impressed  voltage  help  to  determine 
the  rupture  voltage.  When  the  applied  voltage  is  alternating,  the  wave  shape  and  fre- 
quency of  the  voltage  affect  the  dielectric  strength.  Because  of  these  variables,  the  dielec- 
tric strength  in  volts  per  mil  is  not  a  constant  and  is  usually  expressed  as  a  voltage  range. 

Capacitance  varies  inversely  as  the  plate  separation  or  dielectric  thickness.  The  limiting 
minimum,  thickness  is  the  voltage  that  the  capacitor  is  required  to  withstand.  Therefore, 
the  higher  the  dielectric  strength,  or  breakdown  potential,  of  the  insulating  medium,  the 


VARIABLE  AND  ADJUSTABLE  CAPACITORS      3-55 

thinner  the  dielectric  and  the  larger  the  capacitance  value  for  a  given  electrode  area.  This 
can  best  be  illustrated  by  comparing  ceramic  dielectric  to  the  oxide  film  dielectrics  of 
electrolytic  capacitors.  Most  ceramic  dielectrics  are  limited  to  a  breakdown  potential  of 
less  than  100  volts  per  mil,  whereas  the  oxide  film  thickness  of  an  electrolytic  capacitor  is 
measured  in  microns  and  will  withstand  a  voltage  stress  equivalent  to  a  million  volts  per 
mil  of  dielectric  thickness. 

D-c  Leakage.  Continued  polarization  of  a  capacitor  by  direct  voltage  after  full  electri- 
fication results  in  a  flow  of  current  termed  the  leakage  current.  Though  the  d-c  leakage 
is  negligible  under  most  conditions  of  operation,  it  varies  with  temperature  and  should  be 
taken  into  account  in  circuit  designs,  especially  those  involving  grid  coupling  capacitors 
or  similar  circuits  which  are  affected  by  the  flow  of  conduction  current  in  the  capacitor. 

Dielectric  Polarization  and  Absorption.  Capacitors  with  solid  dielectrics  take  longer  to 
charge  than  would  be  predicted  from  their  theoretical  constants,  owing  to  a  lag  or  delay 
in  polarizing  the  dielectric  medium.  This  is  known  as  dielectric  polarization.  When  a 
capacitor  is  discharged  by  short  circuiting,  all  its  energy  is  not  released  and  it  will  build  up 
a  new  charge  with  time  which  is  known  as  the  residual  effect.  This  characteristic  is  known 
as  dielectric  absorption.  Both  are  detrimental  to  circuits  requiring  rapid  charge  and  dis- 
charge characteristics.  Selection  of  a  dielectric  like  polysty-  _  

rene  is  dictated  in  circuits  where  dielectric  absorption  must  be  " 

held  to  the  absolute  minimum. 

Figure  2  illustrates  schematically  what  takes  place  in  a  ca- 

pacitor  having  d-c  leakage  and  dielectric  absorption.    C  repre-    

sents  the  geometric  capacitance  based  on  a  perfect  dielectric. 

Ci  and  n  represent  the  absorption  effect.    The  pure  conduction    FlG'  2'      cSJSS^Cireuit  ° 

effect  is  represented  by  r* 

Power  Factor,  Q,  or  A-c  Resistance.  When  a  capacitor  is  operated  under  a-c  voltage 
stresses,  the  dielectric  will  heat  up,  depending  upon  the  frequency  of  the  applied  voltage. 
The  a-c  resistance  or  the  power  factor  is  a  measure  of  the  heat  dissipated.  It  is  more  con- 
venient in  working  with  electrical  circuits  to  use  a  ratio  of  the  pure  reactance  to  the 
effective  resistance  which  is  termed  the  Q  factor.  The  Q  factor  can  be  employed  for  the 
purpose  of  comparing  various  capacitors  quantitatively  since  it  constitutes  a  figure  of 
merit  for  a  given  design. 

It  is  easily  seen  that,  although  so-called  fixed  capacitors  may  be  made  up  using  any 
one  of  the  many  different  types  of  dielectrics,  the  actual  capacitance  is  a  variable  depend- 
ing upon  the  physical  characteristics  of  the  dielectric,  which  vary  with: 

A.  Voltage,     Capacitance  will  change  with  voltage.    The  voltage  coefficient  will  vary 
with  the  magnitude  of  the  d-c  or  a-c  impressed  voltages. 

B.  Frequency.    The  capacitance  will  change  with  frequency  and  is  a  function  of  the  Q 
or  power  factor  of  the  dielectric. 

C.  Time.    Capacitance  will  change  with  time  because  of  dielectric  aging. 

D.  Temperature.     Capacitance  values  will  change  with  temperature  since  dielectrics 
change  with  temperature  or  have  a  temperature  coefficient. 

These  characteristics  serve  to  demonstrate  the  fallacy  of  considering  a  capacitor  as  an  ideal 
fixed  capacitance. 

Inductance.  In  addition  to  the  limitations  imposed  by  the  dielectric  media,  another 
limitation,  which  results  from  mechanical  design,  is  inductance.  All  capacitors  have  a 
self-resonant  frequency  and  behave  like  a  circuit  involving  series  inductance,  resistance, 
and  capacitance.  Above  the  critical  frequency,  the  reactance  is  inductive  and  not  capaci- 
tive.  In  fact,  capacitors  behave  like  a  complex  impedance  depending  upon  the  operating 
frequency  range. 

18.  VARIABLE  AND  ADJUSTABLE  CAPACITORS 

Capacitors  designed  for  frequent  adjustment  by  an  equipment  operator  are  usually 
termed  variable  or  tuning  capacitors.  Capacitors  of  this  type  most  often  use  an  air  di- 
electric, although  other  dielectrics  are  suitable  for  special  circumstances.  For  example, 
compressed  air  or  nitrogen  dielectrics  reduce  the  size  of  high-voltage  capacitors  in  large 
transmitters.  Mineral  oil,  silicone  fluid,  and  other  liquid  dielectrics  save  space  because 
of  their  higher  dielectric  constant.  In  addition,  such  capacitors  have  a  higher  breakdown 
voltage  for  the  same  interelectrode  spacing. 

Capacitance  adjustment  in  variable  capacitors  is  usually  made  by  varying  the  effective 
plate  area.  The  capacitor  consists  of  two  sets  of  parallel  intermeshed  plates,  one  fixed 
and  one  rotatable  on  a  mounting  shaft.  Rigidity  of  the  capacitor  framework  and  freedom 
from  warpage  of  the  plates  and  the  stator  insulation  are  extremely  important  from  the 


3-56 


RESISTORS,  INDUCTORS,   CAPACITORS 


standpoint  of  circuit  stability.  For  extremely  stable  circuits  in  high-grade  electronic  equip- 
ment, carefully  designed  Invar  steel  frames  and  plates  may  be  used  to  minimize  capacitance 
shifts  with  ambient  temperatures.  More  commonly  frames  and  plates  are  of  aluminum 
or  silver-plated  brass.  In  the  cheapest  broadcast  receivers,  cadmium-plated  steel  has  been 
used.  To  facilitate  tracking  of  circuits  in  broadcast  receivers,  the  outer  rotor  plates  are 
sometimes  slotted  to  permit  small  adjustments  in  the  capacitance-rotation  curve.  The 
supporting  insulation  for  the  capacitor  stator  is  usually  phenolic  or  ceramic,  depending 
upon  circuit  considerations.  Electrically  low-loss,  dimensionally  stable  steatite  or  glass- 
bonded  mica  insulation  is  used  in  capacitors  where  high  Q  and  low  capacitance  drift  are 
important.  In  the  highest  grade  of  precision  laboratory  standard  capacitors  the  insulation 
is  quartz. 

In  certain  highly  accurate  instruments,  the  "standard"  variable  capacitors  consist  of 
two  co-axial  cylinders,  one  fixed  and  one  movable. 

Capacitors  intended  for  relatively  infrequent  adjustment  of  capacitance  are  termed 
"trimmer  or  adjustable"  capacitors.  Most  common  in  broadcast  receivers  is  the  small  mica 
trimmer  capacitor.  Such  a  capacitor  consists  of  a  fixed  and  a  hinged  movable  metal 
electrode  mounted  on  a  phenolic  or  ceramic  base  with  a  mica  spacer  between  the  two  elec- 
trodes. The  movable  leaf  is  raised  or  lowered  by  threading  it  on  a  screw.  These  capacitors 
are  sometimes  called  "book  mica  trimmers." 

In  high-grade  electronic  equipment,  especially  for  frequencies  above  the  standard  broad- 
cast band,  trimmer  capacitors  similar  in  construction  to  intermeshed  plate  variable  capaci- 
tors are  used.  In  many  cases,  they  have  a  shaft  positioning  locking  device  such  as  a 
split  tapered  bushing  and  nut.  Also  found  in  such  equipment  are  adjustable  ceramic 
capacitors.  These  capacitors  consist  of  two  coaxial  half-silvered  ceramic  disks,  one  fixed 
and  one  rotable.  The  ground  unsilvered  faces  are  kept  in  contact  by  spring  pressure. 
The  capacitance  depends  on  the  extent  of  overlap  of  the  silvered  faces.  Through  selection 
of  various  ceramic  compositions  an  opportunity  is  provided  for  some  degree  of  circuit 
temperature  compensation. 

Transmitter  neutralizing  capacitors  are  a  special  design  of  adjustable  capacitors  consist- 
ing of  a  fixed  disk  with  rounded  edges  and  a  similar  coaxial  movable  disk  mounted  on  the 
end  of  a  set  screw.  Capacitance  adjustment  is  by  turning  the  screw  in  and  out  of  a 
threaded  support. 

In  high-powered  aircraft  transmitters,  adjustable  vacuum  capacitors  are  sometimes 
employed.  These  capacitors  have  a  metal  bellows  to  permit  positioning  the  movable 
capacitor  element. 

The  maximum  capacitance  of  variable  capacitors  usually  employed  in  electronic  circuits 
is  from  10  to  530  jujuf .  For  use  as  a  capacitance  standard,  units  are  made  with  capacitances 
up  to  5000  npf.  Trimmer  capacitors  may  have  a  maximum  capacitance  of  5  to  75  pid- 

Since  air  dielectric  capacitors  have  very  little  loss  in  the  dielectric  and  supporting 
insulation,  they  will  show  practically  no  change  in  capacitance  with  frequency.  Accurate 
measurements  of  capacitance  values  at  audio  frequencies  may  therefore  be  depended  upon 
at  radio  frequencies  with  a  good  degree  of  precision. 

EFFECT  OF  STRAY  CAPACITANCE.  Units  of  variable  capacitance  generally  are 
for  relatively  small  values,  and  in  a  circuit,  particularly  at  high  frequencies,  the  capacitance 
effects  of  the  various  other  parts  of  a  circuit  will  be  appreciable  as  compared  with  the 
capacitor  unit.  Every  part  of  the  apparatus  has  capacitance  to  other  parts,  and  these 
small  stray  capacitances  may  be  appreciable.  The  stray  capacitances  are  particularly 
objectionable  because  they  vary  when  parts  of  the  circuit  or  conductors  near  by  are  moved, 
such  as  the  hand  or  body  of  the  operator.  The  disturbing  effects  may  be  minimized  in 
practice  as  follows:  (1)  by  keeping  the  capacitor  a  considerable  distance  away  from  con- 
ducting or  dielectric  masses;  (2)  by  shielding  the  capacitor,  that  is,  surrounding  the  whole 

capacitor  by  a  metal  covering  connected  to 
one  of  the  sets  of  plates;  (3)  by  using  a  capaci- 
tor of  sufficiently  large  capacitance  so  that 
stray  capacitances  are  negligible  in  compari- 
son. The  first  two  of  these  methods  reduce 
only  the  stray  capacitances  of  the  capacitor 
itself  to  other  parts  of  the  circuit  and  to  ex- 
ternal moving  bodies.  Although  the  third 
procedure  is  workable  for  the  lower  radio  fre- 
quencies, it  fails  at  the  very  high  and  ultra- 
high  frequencies,  and  an  entirely  different 
approach  to  the  problem  has  been  necessary. 
Karplus  has  described  a  variable  capacitor  design  which  includes  the  circuit  inductance. 
In  this  design,  wide  tuning  range  with  simultaneous  change  of  lumped  capacitance  and 


FIG.  3.     Semi-butterfly  Circuit 


VARIABLE  AND  ADJUSTABLE  CAPACITORS      3-57 

inductance  is  obtained  by  rotation  of  a  member  that  does  not  require  any  electrical  con- 
nections. This  arrangement  is  called  the  butterfly  or  semi-butterfly  circuit,  depending 
upon  the  configuration  of  the  variable  capacitor  design.  Figure  3  shows  such  a  circuit. 

Maximum  and  minimum  capacitance  of  a  butterfly  circuit  can  be  computed  as  in  a 
variable  air  capacitor,  but,  owing  to  the  butterfly  shape,  capacitance  ratios  are  considerably 
less  than  in  well-designed  tuning  capacitors. 

CAPACITANCE  FORMULA.  The  general  expression  for  the  capacitance  of  a  multi- 
plate  capacitor  is  (in  micro-microfarads) 

C  =  0.08852S:  (Ar~  1)S  (1) 

in  which  S  is  the  area  in  square  centimeters  of  a  moving  plate  overlapping  a  fixed  plate; 
T  is  the  separation  of  plates  in  centimeters;  K  is  the  dielectric  constant  (for  the  air  capaci- 
tor, K  =  1)  ;  JV  is  the  total  number  of  similar  plates  (fixed  plus  movable)  ,  alternate  plates 
being  connected  in  parallel.  In  eq.  (1)  no  correction  is  made  for  the  curving  of  the  lines 
of  force  at  the  edges  of  the  plates;  this  effect  is  negligible  for  most  capacitors,  when  T  is 
very  small  compared  with  S. 

The  numerous  applications  of  variable  capacitors  in  radio  circuits  place  different  require- 
ments upon  the  capacitor  in  its  characteristic  variation  with  setting.  For  laboratory  use  a 
capacitor  is  usually  designed  with  semicircular  plates.  With  this  form  the  capacitance 
increases  linearly  with  the  angular  displacement  of  the  movable  plates.  In  other  radio- 
circuit  applications  it  is  convenient  to  have  the  capacitance  vary  with  the  setting  to  some 
other  power  than  the  first,  and  these  forms  are  noted  below. 

Plate  Form,  Semicircular.  In  a  variable  capacitor  of  the  semicircular  type  the  effective 
area  of  the  plates  is  changed  by  rotating  the  movable  plates.  Throughout  the  entire  plate 
rotation  the  capacitance  is  proportional  to  the  setting,  provided  that  the  capacitor  is  well 
constructed  and  the  distance  between  the  two  sets  of  plates  is  not  affected  by  rotation  of 
the  movable  set.  When  the  plates  are  entirely  unmeshed  (no  overlap),  measurement 
usually  will  show  that  there  is  an  appreciable  capacitance  between  the  capacitor  terminals. 
For  a  5000-/zMf  capacitor  this  may  be  as  large  as  75  jujuf  .  This  value  is  practically  unaffected 
by  the  position  of  the  movable  plates  on  the  rotor,  and  it  represents  the  capacitance  be- 
tween the  insulated  binding  posts  and  the  small  capacitance  formed  across  the  insulators 
which  separate  the  plates.  The  capacitance  of  a  variable  capacitor  cannot,  therefore, 
be  reduced  to  zero,  and  such  relations  as  are  developed  for  capacitance-rotor  positions  hold 
only  when  the  plates  are  enmeshed  a  considerable  amount.  Throughout  the  range  in 
which  the  capacitance  curve  is  linear  the  capacitance  is  given  by  the  expression 
C  =  ad  •+•  Z>,  in  which  6  is  the  angle  of  rotation  of  one  set  of  plates  with  respect  to  the  other. 

Plate  Form,  Logarithmic.  In  a  special  form  of  wavemeter,  called  the  decremeter,  the 
plates  are  so  formed  as  to  determine  the  logarithmic  decrement  of  a  circuit.  This  is  meas- 
ured by  the  percentage  change  in  capacitance  required  to  reduce  by  a  certain  amount  the 
indication  of  an  instrument  in  the  circuit  at  resonance.  In  order  that  equal  angular  rota- 
tions may  correspond  to  the  same  decrement  at  any  setting  of  the  capacitor,  it  is  necessary 
that  the  percentage  change  in  capacitance  for  a  given  rotation  shall  be  the  same  at  all 
parts  of  the  scale.  In  order  that  each  degree  displacement  give  the  same  percentage 
increase  in  capacitance  the  following  requirement  must  be  satisfied: 

^  =  adO  (2) 

where  a  =  constant  =  percentage  change  of  capacitance  per  scale  division.    Integrating, 

log  C  -  ad  4-  6  (3) 

where  5  =  a  constant 

C  =  ««*+&>  =  Coe06  (4) 

where  Co  =  eb  =  capacitance  when  9  =  0. 
Since  the  area  must  vary  as  the  capacitance: 

A  -  1/2  />  dd  ==  Coe06  (5) 

using  polar  coordinates 


and  _  . 

r  =  V2CQaea6  (7) 

The  last  expression  is  the  polar  equation  of  the  bounding  curve  required  to  give  a  uniform 
decrement  scale. 


3-58 


RESISTORS,   INDUCTORS,   CAPACITORS 


Plate  Form,  Wavelength.  For  certain  applications  it  is  desirable  to  have  the  wave- 
lengths proportional  to  the  setting  of  the  capacitor.  Since  the  wavelength  varies  as  the 
square  root  of  the  capacitance,  this  will  require  that  the  capacitor  plates  be  formed  so 
that  the  capacitance  of  the  circuit  varies  with  the  square  of  the  setting.  In  designing  the 
shape  of  these  plates,  allowance  should  be  made  for  the  stray  capacitance  of  the  circuit. 
For  the  design  of  the  simple  straight-line  wavelength  capacitor  the  following  expressions 

axe  used:  /0, 

C  =  A  =  ae2  (8) 

In  polar  coordinates  the  area 


A  -  V2    V  dd 


(9) 


Differentiating  both  expressions: 


(10) 

which  is  the  polar  equation  of  the  bounding  curve  to  give  the  desired  characteristic. 

Plate  Form,  Frequency.  In  another  type  of  capacitor,  of  much  more  importance  than 
those  mentioned  above,  the  variation  of  capacitor  setting  is  directly  proportional  to  the 
frequency  for  which  the  total  circuit  is  tuning.  The  importance  of  this  type  follows  from 
certain  considerations  in  radio  communication,  which  require  the  spacing  of  broadcasting 
stations  from  each  other  by  equal  increments  in  frequency.  It  is  this  condition  which 
makes  highly  desirable  a  capacitor  so  designed  that  equal  increments  in  capacitor  setting 
advance  by  equal  increments  the  frequency  to  which  the  associated  circuit  is  tuned. 
Such  a  capacitor  will  tune  for  the  various  broadcasting  stations  at  equally  spaced  points 
in  the  capacitor  dial. 

In  making  this  special  form  of  variable  capacitor  the  shapes  of  either  rotor  or  stator 
plates  may  be  adapted  to  perform  the  required  area-setting  variation.  It  is  usually  more 
convenient  to  use  ordinary  semicircular  stator  plates  and  form  the  rotor  plates  to  the 
required  shape.  On  this  basis  Forbes  has  shown  that  the  radius  vector  to  the  edge  of  the 
rotor  plate  must  satisfy  the  relation: 


(ID 


in  which  D  =  l/(2irVL);  L  =  inductance  of  the  circuit  in  henries;  n  =  number  of  di- 
electric spaces;  fc  =  lQ-ll/(36ird) ;  d  =  length  of  air  gap  between  plates;  K  =  (/o°  -  /iso0)/*- 

=  cycles  per  radian  of  capacitor  scale;  /o° 
=  frequency  of  circuit  with  capacitor  set  at 
0  deg;  /iso0  =  frequency  of  circuit  with 
capacitor  set  at  180  deg;  Co  =  total  capaci- 
tance of  circuit  when  capacitor  is  set  at  0 
cleg — this  includes  stray  circuit  capacitance 
as  well  as  that  of  the  zero  setting  of  the  ca- 
pacitor in  farads;  9  =  angle  of  rotation  of 
the  plates,  radians;  and  n  =  radius  of  the 
cut-out  of  stator  plates  to  accommodate  the 
rotor  shaft.  All  dimensions  in  centimeters. 
The  capacitance  of  such  a  capacitor,  for  any 
angular  position  6  of  the  rotor,  is 

D 


500 
f 

1  400 
£ 


20° 


40         60         80 
Sea  e  Reading.  Divisions 


100 


100  _         _ 

'~  (12) 

Typical  variations  of  capacitance  with 
setting  are  shown  in  Fig.  4,  which  represents 
FIG.  4.    Capacitance  Calibrations  for  Three  Typi-    the  construction  practice  of  one  manufac- 
^^/iS^^^rSl!1^!  turer  of  variable  capacitors.     The  curves 
Straight-line  Frequency  shown  indicate  the  value  of  capacitance  ob- 

tained  for   various  settings   of  the   three 

general  types  of  capacitors,  straight-line  capacitance  (SLC),  straight-line  wavelength 
{&LW),  and  straight-line  frequency  (SLF). 


VARIABLE  AND  ADJUSTABLE  CAPACITORS 


3-59 


Plate  Form,  Special  Designs.  Special  considerations  must  be  given  to  the  design  of 
variable  air  condensers  when  two  or  more  units  are  connected  together  for  operation  on  a 
common  shaft.  This  grouping  of  sections  into  "gangs"  sometimes  places  five  condensers 
on  a  single  control,  and  these  may  perform  important  functions  in  as  many  individual 
circuits.  An  illustration  of  this  construction  occurs  in  a  radio  receiver  in  which  there  are 
two  identical  units  for  tuning  signal  frequency  circuits,  one  unit  with  specially  shaped 
plates  to  tune  the  oscillator  circuit  and  two  smaller  units  to  tune  the  short-wave  circuits. 
In  order  to  integrate  the  actions  of  all  these  condensers  in  the  above  superheterodyne 
radio  receiver,  where  the  single  control  is  particularly  desirable,  various  methods  of  design 
have  been  developed.  E.  D.  Koepping  has  described  the  general  principles  involved, 
D.  F.  McNamee  has  presented  a  graphical  solution  of  the  design  problem,  and  H.  Schwartz- 
mann  has  shown  an  analytical  solution  of  the  problem  (see  Bibliography,  p,  3-73). 

PLATE  SPACING  OF  AIR  CAPACITORS.  Between  two  oppositely  charged  paraUel 
plates  of  infinite  extent  the  field  is  uniform.  It  is  known,  however,  that  the  field  between 
two  finite  plates  is  not  uniform,  tending  to  greater  voltage  gradients  at  the  edges.  Stress 
at  a  point  in  a  dielectric  is  determined  by  the  gradient  at  that  point,  and  stresses  above 
certain  critical  values  lead  to  the  formation  of  corona  and  sparkover.  This  may  be  pictured 
as  follows,  with  plate  spacing  =  /S  and  thickness  as  T.  For  large  values  of  S/T,  the  radius 
of  the  plate  edge  is  small  and  a  high  gradient  exists.  As  the  plate  thickness  is  increased, 
keeping  plate  center-to-center  spacing  and  applied  voltage  constant,  the  radius  of  edge 
curvature  is  increased.  This  decreases  the  edge  gradient.  Inside  the  capacitor,  however, 
the  gradient  increases  because  of  the  reduced  spacing.  It  has  been  suggested  by  Ekstrand 
that  the  best  value  of  S/T  may  be  between  2  and  3.  Investigation  seems  to  corroborate 
Ekstrand  figures  by  fixing  the  optimum  S/T  at  2.77. 

Plate  Dimensions  and  Voltages  of  Capacitors  at  Various  Frequencies 


T 

in. 

S 
in. 

S/T 

Sparkover  Voltage 

Maximum  Gradient 

kv 
©  60  cy. 

kv 
©  700  kc 

kv 
©  1500  kc 

kv/in. 
@  60  cy. 

kv/in. 
@  700  kc 

kv/in. 
©  1500  kc 

0.128 
0.04 
0.064 

0.218 
0.192 
0.719 

1.705 
4.8 
11.24 

14 
8.4 
24 

13.5 
7.59 

14.28 

13.7 
6.82 
11.7 

89.1 
80.2 
82.0 

85.8 
72.5 
48.8 

87.1 
75.1 
40.0 

It  has  become  a  habit  to  consider  breakdown  as  coincidental  with  sparkover.  Actually, 
breakdown  has  occurred  at  the  first  sign  of  corona,  and  corona  may  become  evident  at  a 
considerably  lower  voltage  than  sparkover.  When  a  conductor  is  raised  beyond  a  certain 
critical  potential,  'the  air  adjacent  to  it  becomes  ionized,  forming  corona.  The  ionized  air, 


2000 
4000 
6500 
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)0          500          700          900       1100        1300        1500       1700       1SOO       2100       2300 
D-c  breakdown  voltage 

FIG.  5.     Breakdown  Voltage  vs  Altitude  (Atmospheric  Pressure) 


3-60 


RESISTORS,   INDUCTORS,    CAPACITORS 


being  itself  a  conductor,  can  be  considered  to  increase  the  dielectric  losses.  If  the  gradient 
is  further  increased,  the  conditions  become  unstable  and  sparkover  occurs. 

Not  only  are  capacitors  used  in  transmitters  and  electronic  heating  devices  subjected  to 
high  voltages,  but  they  also  must  carry  considerable  amounts  of  power.  The  power  input 
to  a  capacitor  is  measured  by  the  energy  stored  in  it  multiplied  by  the  number  of  charges 
and  discharges  per  second. 

Transmitter  capacitors  used  in  aircraft  electronic  equipment  must  have  their  plates 
spaced  properly  to  withstand  voltages  encountered  at  maximum  flying  altitudes.  The 
curves  of  Fig.  5  indicate  breakdown  voltage  for  various  plate  spacings  at  various  altitudes. 


19.  IMPREGNATED-PAPER  CAPACITORS 

Impregnated-paper  capacitors  are  the  most  efficient  of  all  types  because  of  their 
flexibility  in  size,  shape,  and  rating.  They  cover  an  extremely  wide  range  in  size  from  the 
small  toothpick  varieties  found  in  hearing  aids  to  large  welded  case  blocks.  Flexibility 
in  voltage  ratings  is  second  only  to  size  values  with  a  range  from  1  volt  to  200,000  volts. 
Capacitance  values  are  common  in  a  range  from  lOOju/xf  to  200^f  in  a  single  container. 

Electronic  equipment  designers  are  primarily  concerned  with  d-c  application  of  impreg- 
nated-paper capacitors. 

D-c  service  includes  such  applications  as  rectifier  niters,  energy  storage,  arc  suppression, 
and  by-passing  for  electron-tube  and  circuit  elements.  D-c  ratings  provide  for  small  a-c 
components  where  their  heating  effects  are  negligible.  Experience  indicates  that  the  a-c 


Edges  of  foil  sweflged 
both  ends 


Thined  copper  tabs 


Inductive  and  Non-inductive  Capacitor  Windings 


components  should  not  exceed  20  per  cent  of  the  d-c  value  at  a  frequency  of  60  cycles,  or 
15  per  cent  for  120  cycles,  or  1  per  cent  for  10,000  cycles,  and  for  higher  frequencies  the 
allowable  magnitude  is  determined  strictly  on  thermal  evaluation  of  the  high  frequency 
component. 

Other  special  electronic  applications  require  rating  of  the  capacitor  on  an  a-c  rather  than 
a  d-c  basis,  such  as  tuned  niters  and  pulse  networks. 

A-c  versus  D-c  Ratings.  It  is  possible  to  rate  capacitors  designed  for  a-c  service  in 
terms  of  d-c  voltages,  but  it  is  not  practical  to  rate  d-c  capacitors  in  terms  of  a-c  voltage 
ratings  because  of  the  difference  in  design  considerations  in  the  d-c  rated  capacitors  in 
comparison  to  those  rated  for  alternating  current. 

Some  of  these  design  considerations  are: 

1.  D-c  voltage  ratings  depend  essentially  on  dielectric  stress. 

2.  A-c  voltage  ratings  depend  not  only  on  dielectric  stress  but  also  on  the  operating 
frequency  and  power  factor.    Frequency  and  power  factor  determine  the  internal  heating, 
which  must  be  kept  within  limits  determined  by  the  radiating  surface  of  the  container. 
For  small  a-c  capacitors,  the  ratings  are  determined  from  dielectric  breakdown  voltage 
considerations.    In  the  larger  a-c  voltage  ratings,  the  prime  design  consideration  is  heating. 

In  general,  the  following  tabulation  represents  the  nearest  standard  d-c  rating  corre- 
sponding to  the  several  standard  a-c  voltage  ratings: 


A-c  VOLTS 
110 
220 
330 
440 
550 
660 


D-c  VOLTS 

200 

400 

600 
1000 
1500 
2000 


Chlorinated  diphenyl  impregnated  capacitors  designed  for  a-c  service  cannot  be  used 
on  direct  current  unless  the  impregnant  has  been  chemically  treated  with  inhibitors  or 


IMPREGNATED-PAPER  CAPACITORS 


3-61 


stabilizers  to  prevent  deterioration  of  the  impregnant  from  the  combined  influence  of  the 
d-c  field  and  high  temperature. 

Construction  of  Winding.  Impregnated-paper  capacitors  are  made  in  a  roll  construc- 
tion, consisting  of  two  metallic  foils  separated  by  two  or  more  sheets  of  impregnated  kraf t 
tissue.  In  the  roll  construction  the  resultant  capacitance  is  twice  that  obtained  with  a 
parallel-plate  construction  since  both  sides  of 
the  foils  are  active.  The  capacitor  winding 
may  be  round  or  flat,  depending  upon  mechani- 
cal considerations  of  housing. 

A  further  consideration  in  roll  construction 
is  whether  the  foils  are  of  the  "buried"  type 
with  tabs  for  contact  members  or  of  the  ex- 
tended foil  construction.  See  Fig.  6(a)  and  (6). 
The  extended  foil  construction  gives  the  low- 
est value  of  self-inductance  since  all  the  turns 
are  bonded  together,  the  construction  ap- 
proaching that  of  a  stacked  parallel-plate 
capacitor.  The  "buried"  foil  winding  ap- 
proaches the  "extended"  foil  if  the  tabs  are 
inserted  at  the  center  of  the  winding  within 
one  turn  of  each  other.  Figure  7  shows  a  com- 
parison of  the  impedance  for  "extended"  vs. 
"buried"  foil  constructions  for  a  frequency 
range  of  60-5000  cycles. 

Impregnation  of  Winding.  The  impregnants  most  commonly  used  with  kraft  paper 
are  microcrystalline  hydrocarbon  waxes,  chlorinated  waxes  and  oils,  castor  oil,  mineral 
oil,  and  plastic  compounds.  These  various  impregnants  offer  a  wide  range  of  characteristics 
with  temperature.  Figure  8  shows  six  of  the  more  common  wax  and  oil  impregnants  for  a 
temperature  range  of  —40  to  100  deg  cent.  These  data  were  taken  for  average  production 
capacitors  and  do  not  represent  minimum  requirements.  Minimum  requirements  are 
shown  in  the  accompanying  Table  1. 

Voltage  Rating  of  D-c  Capacitors  with  Temperature  and  Service  Conditions.  All 
capacitors  are  affected  by  temperature,  voltage  stress,  and  time. 

Table  1.     Minimum  Impregnant  Requirements 


Power  factor  tn  per  ceTjt 

i-»  »-«  M  M  OJ  O>  -N  -1 

010  01  O  01  o  01  o  tri  oc 

/ 

/ 

/ 

Inductive,  tat! 
center  of  wir 

s  fn 
ding    j 

/ 

A 

In 

e 

du 
nc 

"ctve,  tabs  at 
of  windings^/ 

/ 

\ 

^^ 

\ 

^ 

***• 

*** 

*** 

"• 

.—-- 

•inductiv 

~*^ 

e  win 

ding 

—  "  Non 

)0                1000              2000     3000         50C 

Frequency  In  cycJes 

FIG.  7.  f  Comparative  Power  Factors  for  Induc- 
tive and  Non-inductive  Windings 


Castor 
Oil 

Mineral 
Oil 

Chlorinated 
Diphenyl 

Halowax 

Megohms  times  microfarads: 
At  25°  C     

500 

2000 

1500 

2000 

At  high  test  temperature  *                                   

5 

20 

15 

100 

Insulation  resistance  in  megohms: 
At  25°  C            

1500 

6000 

4500 

6000 

At  high  test  temperature                                                .  • 

150 

600 

450 

1000 

Capacitance  change  in  per  cent  at  low  ambient  test 
temperature  from  value  at  25°  C  t  

-30 

-15 

-30 

-10 

*  85°  C  for  all  impregnants  except  Halowax,  which  is  measured  at  65°  C. 
f  Lowest  temperature  —40°  C  except  for  Halowax,  which  is  —20°  C. 

It  has  been  standard  practice  to  rate  d-c  capacitors  at  a  40  deg  cent  ambient  temperature 
on  the  basis  that  such  a  rating  would  provide  sufficient  factor  of  safety  to  withstand  a  life 
test  of  1000  hours  at  twice  the  rated  voltage  at  this  temperature,  which  is  equivalent  to 
approximately  1  year  of  normal  service  conditions. 

Design  trends  for  electronic  equipment  toward  smaller  and  smaller  physical  volume  have 
resulted  in  ambient  temperatures  considerably  in  excess  of  the  40  deg  cent  value  considered 
standard  before  World  War  II. 

It  has  not  been  generally  understood  that  increasing  the  ambient  temperature  above 
40  deg  cent  required  voltage  derating  for  an  equivalent  life  expectancy  at  the  higher  tem- 
peratures, and  that  the  derating  factors  are  a  function  of  capacitor  size.  Capacitor  size 
can  best  be  evaluated  for  d-c  ratings  in  terms  of  energy  content  in  watt-seconds. 

Watt-second  =  */2  CE2 

where  C  =  capacitance  in  microfarads  and  E  —  d-c  voltage  in  Mlovolts. 

This  was  a  problem  given  joint  industry  and  government  appraisal  during  the  war,  and 
the  findings  resulted  in  a  table  of  derating  factors  as  a  function  of  temperature  and  capaci- 


3-62  RESISTORS,   INDUCTORS,   CAPACITORS 


|      1  —  h- 

^M^ 

*    ?| 

N 

v~ 

=50, 

noc 

=4^ 

}3 

1    ' 

"N- 

"3 

N 

-2 

?'C 

00 

r 

\ 

10,000  -S 

1 

vv 

5,000  |«5 

1 

2,000'|^ 

\ 

7 

Mi 

ier£ 

1  0 

i 

\ 

1,000  1  1 

fnj= 

V 

S 

o 

2 

00- 

50 

\J 

°1 

& 

100 

— 

1 

9^ 

Sn 

-— 

Q.  Q 

I"*TH 

'1.              In        !       '    j 

|-l-2 

1-2 
8-4 


•50,00 


-+—F 


2,000 


10,000  rt 
5,000  11 


S-g 


1,000  gx 

500 

200 


ng 
So 


tfl-rs- 

«4-2 
^o     0 

S-2 

g     ^ 

•^^ 

—  « 

^ 

«5^ 

Insulation  resistance  «-» 
Megohms  X  microfarads 

^ 

"""v. 

^—4 

ro 
p. 
o 

S 

o 

a 
S 

IB 

*c  2 

1 

-—  - 

M,!C 

lydr 

roc 
oc< 

••^i 
7bq 

^ 

a!ltn€ 
n  wa 

u 

10,00 

^ 

5,000 

N 

N 

2,( 
1,C 

50 

20 

10 

300 
)00 
0 

0 
0 

S 

^ 

==5 

=1 

^-40     -20        0    -t-20 +25+40    +60    -1-80     +100 
Temperature  in  degrees  caniigrade 


S,  +2 

|    o 

-1— 

|-10 
8-14 

1 

1 

o  c 

«5 

c2 

F 

^ 

ss' 

0,( 

00 

10 

,00 

ion  resistance  ^ 
5  X  microfarads 

Cas 

tor 

5,000 

oil 

2,C 

l.C 

)00 
00 
0 

3 

^ 

50 

20 
Oy? 

3   0 

«M 

\. 

\ 

s 

= 
s^ 

-1C 
50 

/ 

"1 

c^- 

-^~ 

i  r-- 

>^ 

S.+4 

Per  cent  capacitance  cb 
>er  cent  power  factor  -  1000  oo  ,  ,  , 
oi-»tow  o>itoo 

"i  n 

00- 

!—  ' 

H 

DlU 

lin 

itec 
nib' 

1  Wt, 

tor 

0, 

)OC 

1C 
5,C 

I/ 

50 

20 
10 

,00 
)00 
)00 
)00 
0 

0 
0 

ation  resistance  o 
ms  X  microfarads 

< 

s, 

^ 

\ 

s 

il 

S 

—  ». 

**^ 

•H  — 

1-4° 
S      8 

(^ 

Insulation  resistance  o 
Megohms  X  microfarads 

/* 

sr* 

p,boc 

r~  2 
\ 

o,c 

00 

10,00 

"5_ifi 

—  / 

ite< 
tnh 

\ 

1   d  p 
ibito 

Percentcapa 
ower  factor  -  60  ^  T  I  i 
oo  4,  a.  gg^{ 

y 

^ 

irin 

JlUS 

_^ 

/ 

Jhlc 
I 

henyl 

r 

S 

5,C 
2,C 

1,C 
50 

\ 

)00 
>00 
00 
0 

\ 

\ 

* 

K 

0- 
)0- 

^ 

R 

/ 

-2 

-/L 

^ 

Si 

MM" 

^ 

-i 

=>-  = 

^z= 

*•** 

=±: 

-40    -20        0    +20+25+40     +60    +80    +100 
Temperalore  in  degrees  centigrade 


Typical  Electrical  vs  Temperature  Characteristics  of  Paper-dielectric  Capacitors  with.  Various 
Impregnations  (Courtesy  of  Solar  Mfg.  Corp.) 


IMPREGNATED-PAPER  CAPACITORS 


3-63 


tor  size.  These  data  are  shown  in  Fig.  9  and  are  extracted  from  data  obtained  in  Proposed 
Joint  Army-Navy  Specification  JAN-C-25  dated  January  22,  1945,  and  published  in 
Communications  for  August  1947. 

These  data  show  that  a  capacitor  in  the  0.5  watt-second  class  and  rated  for  a  given 
voltage  at  40  deg  cent  must  be  voltage-derated  to  95  per  cent  of  its  40  deg  cent  rating  at 
85  deg  cent  or  60  per  cent  at  105  deg  cent.  Generally  speaking,  the  life  expectancy  of  a 


Capacitance                                Rated                                Watt  - 
/Xf       A-W-S                                  volts                                     sec 

Permissible  volts 
-Per  cent               at- 

o.oi- 

-103                                                                                    | 

55  C 

65  C 

75  C 

850 

Q. 

0.02- 

-203 

-100,000 

-100  o 

-75,000 

I         ° 

—  50  000 

,, 

85 

80 

65 

45 

0.05- 

-503 

-40,000 

-40.0^ 

-30,000 

-30.0 

-25,000 

o.i- 

=104 

-20,000 

"20.0^ 

-15,000 

g 

-12,500 

0 

0.25~ 

"254 

-10,000 

-10.0 

-7,500 

-6,000 

—  c   QQQ 

90 

85 

70 

55 

0.50- 

-504 

-4,000 

-4,0 

-3,000 

-3.0 

-2,500 

1.0Z 

=105 

-2,000                ^^ 

-2.0  ~ 

-1,500    ^.^"^ 

1 

2,0- 

-205                        ^> 

^Cooo 

Z1.0 

A.  r\ 

"^--^^ 

-600 

95 

85 

75 

65 

4.0* 

•*"405 

-400 

-0.4 

6.0- 

-605 

-0^ 

8.0- 

-805 

-250 

10.0- 

:106 

-200 

-0.2 

12.0- 

-126 

15.0- 

-156 

T-l 

20.0- 

-206 

-100 

=  °'10     | 

30.0- 

-306 

~                         <5 

40.0- 

-406 

-0.05 

50.0- 

-506 

95 

90 

80 

70 

100- 

FIG.  9.     Ratings  of  D-c  Capacitors 

capacitor  without  derating  is  halved  for  each  10  deg  cent  rise  in  temperature.  For  d-c 
voltage  considerations,  at  a  fixed  temperature  of  40  deg  cent,  the  life  expectancy  is  in- 
versely proportional  to  the  fifth  power  of  the  voltage. 

D-c  voltage  ratings  are  not  fixed  values,  but  for  a  given  insulation  thickness  they  may- 
be made  variable  depending  upon  the  duty  cycle  and  circuit  conditions.  A  capacitor  rated 
at  1000  volts  d-c  at  40  deg  cent  for  continuous  duty  in  a  power-supply  filter  could  be  used 
in  a  photoflash  circuit  at  a  much  higher  voltage,  such  as  2000  volts  d-c,  and  still  have 
acceptable  performance  because  of  the  lighter  duty  cycle.  Life  expectancy  might  increase 
to  100,000  flashes  of  the  photoflash  equipment. 

This  indicates  that  considerable  flexibility  may  be  used  in  rating  capacitors  provided 
that  all  the  design  requirements,  operating  conditions,  and  duty  cycle  are  known.  To 
illustrate,  a  10-^f  capacitor  winding  using  three  sheets  of  0.0004  kraft  tissue  and  impreg- 
nated in  mineral  oil  would  be  normally  rated  at  1000  volts  d-c  at  40  deg  cent  continuous 
duty,  at  1200  volts  for  an  intermittent  duty  cycle  of  about  50  per  cent,  at  1500  volts  inter- 
mittent duty  for  welder  applications,  and  2000  volts  under  a  photoflash  duty  cycle.  The 
d-c  rating  would  have  to  be  reduced  to  600  volts  for  long-time  (15  years)  life  expectancy. 

Factors  that  affect  capacitor  ratings: 

1.  Microfarad  value  and  tolerance. 

2.  Duty  cycle,  continuous  or  intermittent. 


3-64  RESISTORS,   INDUCTORS,   CAPACITORS 

3.  Ambient  temperature  range. 

4.  Ripple  voltage,  magnitude,  and  frequency. 

5.  Abnormal  circuit  voltages,  such  as  no  load  voltage  and  peak  charging  voltage. 

6.  Discharge  current,  and  nature  of  discharge,  whether  oscillatory,  and,  if  so,  whether 
critically  damped. 

It  is  well  to  remember  that  these  factors  that  govern  ratings  are  based  on  hermetically 
sealed  capacitors  which  have  been  carefully  dried,  impregnated,  and  sealed.  They  do  not 
apply  to  other  constructions  where  life  is  limited  by  the  vagaries  introduced  by  moisture 
absorption. 

METALLIZED  PAPER  CAPACITORS.  The  latest  addition  to  the  family  of  impreg- 
nated-paper capacitors  is  the  MP  type,  in  which  the  capacitor  electrodes  are  deposited 
on  the  paper  dielectric  in  very  thin  films,  having  a  thickness  range  between  25  and  100 
millimicrons.  The  thin  film  contributes  the  property  of  "self-healing"  to  capacitors,  per- 
mitting the  use  of  a  single  sheet  of  dielectric,  which  is  not  possible  with  conventional  im- 
pregnated kraft  paper  designs.  The  combination  of  the  extremely  thin  metallic  film  elec- 
trode and  a  single  sheet  of  dielectric  affords  extremely  compact  designs  for  voltage  ratings 
below  200  volts  d-c  or  for  150  volts  a-c. 

Voltage  ratings  exceeding  200  volts  employ  a  multiple-layer  or  interleaved  paper  di- 
electric of  conventional  construction.  The  volume  saving  is  not  as  great  as  for  the  single- 
layer  construction  but  is  still  considerable. 

A  new  concept  in  capacitor  rating  is  involved  with  MP  capacitors,  namely  sparking 

voltage;  this  is  defined  as  the  lowest  applied  voltage 

Table  2.  D-c  Voltage  Ratings  that  ^  cause  continuous  "self-healing"  action  to 

take  place. 


D-c  Working 

Voltage 

25°  C 


200 
400 
600 


1  Minute 

Flash  Test 

25°  C 


300 
600 
900 


Sparking 

Voltage 

25°  C 


400 
900 
1350 


MP  capacitors  designed  so  that  the  maximum 
surge  voltage  encountered  in  service  at  the  highest 
operating  temperature  does  not  exceed  the  sparking 
voltage  are  usable  in  all  kinds  of  circuits  without 
fear  of  their  causing  spurious  noise.  In  the  event 
of  a  transient  voltage  which  would  cause  failure  of 
a  conventional  capacitor  type,  there  will  be  only  a 
momentary  arc  discharge  followed  by  the  self-healing  mechanism. 

Insulation  Resistance.  The  minimum  insulation  resistance  of  single-layer  lacquered 
metallized  paper  capacitors  will  exceed  500  megohm  microfarads  or  2000  megohms  at  25 
deg  cent.  Interleaved  unlacquered  metallized  paper  capacitors  will  have  a  minimum 
insulation  resistance  of  1000  megohm  microfarads  or  6000  megohms  at  25  deg  cent,  which 
compares  favorably  with  conventional  capacitor  designs.  The  change  in  insulation  resist- 
ance with  temperature  in  metallised  paper  capacitors  is  similar  to  that  of  conventional 
mineral  oil  impregnated  kraft  paper  designs,  or  there  is  approximately  a  50  per  cent  de- 
crease in  insulation  resistance  for  every  10  deg  cent  rise  above  25  deg  cent. 
Commercial  Specification  References. 

Joint   Army-Navy   Specification  JAN-C-25,    Capacitors,   Direct   Current,  Paper   Dielectric,    Fixed 

(Hermetically  Sealed  in  Metallic  Cases) . 

Joint  Army-Navy  Specification  JAN-C-91,  Capacitors,  Paper  Dielectric,  Fixed  (Non-magnetic  Cases). 
RMA  Standards  Proposal  159. 

20,  MICA  CAPACITORS 

Mica  capacitors  are  useful  in  electronic  circuits  because  of  their  low  a-c  losses  and  their 
high  electrical  stability  over  a  wide  temperature  range.  These  characteristics,  along  with 
the  fact  that  they  are  constructed  to  very  close  capacitance  tolerances,  make  them  ideally 
suited  for  use  in  frequency-determining  circuits. 

The  word  "mica"  is  derived  from  the  Latin  "micare"  meaning  to  sparkle.  It  is  a  group 
name  for  a  number  of  aluminum  silicate  minerals  which  are  characterized  by  the  properties 
of  high  reflection  and  a  basal  cleavage  so  perfect  that  they  may  be  split  in  laminae  of  the 
order  of  0.0005  in.  thick. 

Of  the  eight  distinct  species  of  mica  recognized  by  mineralogists,  muscovite  is  the  most 
important  as  far  as  mica  capacitor  manufacture  is  concerned.  Muscovite  is  virtually 
unaffected  by  weathering,  is  not  porous,  is  not  decomposed  by  acids,  and  is  negligibly 
affected  by  moisture.  It  will  withstand  relatively  high  voltage  gradients.  Voltage  tests 
made  with  spherical  electrodes  show  that  films  1/iooo  in.  thick  frequently  withstand  5000 
volts  d-c  with  no  puncture.  Muscovite,  because  of  its  extremely  low  power  factor,  in 
addition  to  its  other  desirable  properties  described  previously,  is  an  ideal  dielectric  for  use 
in  capacitor  manufacture. 

Both  micas  and  foils  or  silvered  electrode  patterns  must  be  precision  outlined.    Micas 


Style  20  Style  25 

' 


Style  56       Style  60         ®tyl8  Sioos 
_      *B.OOJ  _  P«0.14^0>000 


2.375 

.O 


All  dimensions  in  inches     °-150  min' 
Style  65 


••l-fj-max.  > 


ffi~nrj 


10  32  thread  . 

All  dimensions  in  inches  °-180  min 
Style  70 


All  dimensions  in  inches  \T-JI 

-rdia 


Style  75 

When  spark  gap  Js  used 
,— capacitor  will  be  this  shape 


«*  «" 


All  dimensions  in  Inches  \T-O 

Style  80 

When  spark  gap  Is  used 
capacitor  will  be  this  shape 


All  dimensfons  in  inches 


Style  85    • 

When  spark  gap  5s  used 
\capacitor  will  be  this  shape 


^f 


All  dimensions  in  inches 

Style  90 


.  When  spark  gap  Is  used 
10±  ie ^capacitor  will  be  this  shape 

4**  |       H*^ 


All  dimensions  in  inches 

Style  95 


FIG.  10.     Standard  Outline  Dimensions 

3-65 


3-66 


RESISTORS,  INDUCTORS,   CAPACITORS 


for 


are  usually  cut  to  within  plus  or  minus  0.001  in.  of  specified  lengths  and  widths  and  foils 
almost  as  accurately. 

Sealing  binders  as  well  as  metal  clamps  provide  permanent  positioning  of  the  mica  stack. 

This  accuracy  and  permanency  provide  the  sta- 
bility of  characteristics  with  respect  to  aging,  fre- 
quency, and  temperature  which  recommend  mica 
capacitors  for  use  in  frequency-determining  circuits 
or  circuits  that  control  reactance  and  phase  and  in 
precision  measuring  equipment. 

Table  4.     B-c  Voltage  or  Peak  "Working  Voltage 
Ratings  for  the  Several  Case  Types 


Table    3.     Capacitance    Range 
the  Several  Case  Types 


Case  Type 

Prom 

To 

20 

5  wrf 

1  ,  000  md 

25 

5 

1,500 

30 

470 

10,000 

35 

3,300 

10,000 

40 

100 

10,000 

45 

17 

10,000 

50 

2,000 

27,000 

55  and  56 

22 

30,000 

60  and  61 

100 

47,000 

65 

47 

100,000 

70 

47 

100,000 

75 

47 

100,000 

80 

47 

100,000 

85 

47 

100,000 

90 

100 

100,000 

95 

100 

10,000 

Note:  Capacitance  values  for  1000  ntf 
or  less  are  measured  at  or  referred  to 
500  kc/sec.  Capacitance  values  greater 
than  1000  prf  are  measured  at  or  referred 
to  1000  kc/sec. 

Standard     commercial     tolerance    for 


Style 

Voltage  Range 

20,  25,  30,  35 

300  and  500  volts 

40 

300,  500,  1000 

45,  50,  55,  56,  60,  61 

600,  1200,  2500 

65 

250,  500,  1000,  1500,  2000, 

3000 

70 

500,  1000,  1500,  2000,3000 

5000 

75 

1000,  1500,  2000,  3000, 

4000, 

6000 

80 

1500,  2000,  3000,  4000, 

5000, 

6000,  8000,  10,000 

90 

3000,  4000,  5000,  6000, 

8000, 

10,000,  12,000,  15,000, 

20,- 

000 

95 

15,000,  20,000,  25,000,30 

,000, 

35,000 

Mica  dielectric  capacitors  because  of  their  low 


other  tolerances  are  available  as  shown  in 
jrjg>  10. 


for  high  current  circuits. 

The  mechanical  forms  of  mica  dielectric  capacitors 
vary  from  small  phenolic  molded  cases  for  receiver 
applications  to  large  ceramic  insulated  housings  required  by  high-current,  high- voltage- 
transmitting  circuits. 

The  mica  capacitor  is  the  only  one  of  the  many  capacitor  types  that  has  obtained  general 
industry  standardization  for  case  styles  and  electrical  ratings. 

Figure  10 (a)  shows  outline  dimensions  for  molded-case  types,  Styles  20,  25,  30,  35,  40, 
45,  50,  55,  56,  60,  and  61;  Figs.  10(6)  and  (c)  show  outline  dimensions  for  the  potted-case 

types,  Styles  65,  70,  75,  80,  85,  90, 
Table  5.     Classification  and  95. 

Capacitance.  Capacitance  val- 
ues for  mica  capacitors  are  ex- 
pressed in  micro-microfarads. 

Capacitance 
1st  and  2nd  digits  a-nd 

^- ^-  multiplier 


Designation 

Temperature 
Coefficient 
Not  more  than 

Capacitance 
Drift 
Not  more  than 

Class  A 

±1  000  ppm 

±(5%  +  1  wuf) 

Class  B         

±  500  ppm 

±(3%  +  1  ftftf) 

Class  C  

±  200  ppm 

±(0.5%  +  0.5  jajuf) 

Class  I  

(+150          ) 

±(0.3%  +  0.2^/zf) 

Class  D      

(  —      50  ppm  } 

db   1  00  ppm 

±(0.3%  4.  o.l  ji/rf) 

Class  J  

(+100       i 

±(0.  2%  +  0.  2  /i/if) 

Class  E         

I  —     50  ppm  J 

f+ioo        i 

_j_(0  1%  4-  o  1  wzf) 

Class  G 

1  -     20  ppm  j 
/            0           ) 

±(0  1%  +  0  1  wif) 

\  -     50  ppm  J 

White 


Glass 


Tolerance 


FIG.  11.     RMA  Capacitor  Color  Code 
(Proposed) 

Color  Marking  of  Molded  Types. 

Note:  Characters  D,  J,  E,  and  G  require  individual  tests  of  A  convenient  ^ system  of  six-color 
each  capacitor  and  should  be  considered  for  use  only  where  marking  is  being  used  to  identify 
extreme  stability  and  accuracy  are  required.  molded  mica  case  types.  Two 

RMA  color  codes  are  in  use,  one  as 

a  standard  and  the  other  proposed,  but  industry  standardization  is  expected  to 
make  the  old  system,  which  does  not  provide  for  identification  of  the  class  designation, 
obsolete. 

Figure  11  shows  the  arrangement  of  the  two  rows  of  colors,  and  the  significance  of  each 
color  is  shown  in  Table  6. 


CEKAMIC  DIELECTRIC   CAPACITORS 


3-67 


Radio-frequency  Current  Ratings.    The  potted-case  type  of  mica  capacitor  is  intended 
primarily  for  use  in  frequency-determining  circuits  or  those  requiring  the  capacitor  to 

Table  6 


Color 

Numerical 
Significance 

Decimal 
Multiplier 

Capacitance 
Tolerance 

Class 
Designation 

Black   

0 

1 

20% 

A 

Brown.  .  .  . 
Red 

1 
2 

10 
100 

2% 

B 

c 

Orange  
Yellow.... 
Green.    .  .  . 

3 

4 
5 

1,000 
10,000 

3% 
5% 

D 

E 

Blue  

6 

Violet 

7 

Gray 

8 

I 

White  

9 

J 

Gold 

0.  1 

Silver  

0.01 

10% 

15 


handle  appreciable  amounts  of  r-f  current.  Typical  curves  illustrating  the  current- 
carrying  capacity  for  a  frequency  range  of  0.1  to  30  megacycles  of  mica  dielectric 
capacitors  housed  in  potted  ceramic  cases  are  shown  by  Fig.  12. 

It  is  interesting  to  note  the  increasing  current  2o 
loading  with  frequency  for  the  O.OOOl-ju/xf  ca- 
pacitor in  comparison  with  the  decreasing  cur- 
rent rating  with  increasing  frequency  for  ca- 
pacitance values  greater  than  0.001  jujuf,  which 
is  due  to  the  inductance  introduced  in  the 
series-paralleling  of  the  mica  sections  which 
constitute  the  capacitor  stack  for  the  larger 
capacitance  values. 

The  current-carrying  capacity  of  mica  dielec- 
tric capacitors  can  be  materially  increased  by 
immersing  the  foil-mica  stack  in  silicon  oil  and 
removing  the  heat  generated  by  means  of 
cooling  coils. 

Manufacturers'  published  ratings  are  based 
on  an  ambient  temperature  of  40  deg  cent, 
and  the  current  ratings  must  be  derated  for 
higher  ambient  temperatures  as  shown  in 
Table  7. 


20 


=  10 


Table   7.     Current   Deratings   with  Tem- 
perature 


Characteristic 

Temperature 
Range 

Current 
Derating 
Factor 

B,  C 
B,  C 
B,  C 

D,  E,  G 

41  to  50  deg  cent 
51  to  60 
61  to  70 
40  to  70 

0.95 
0.85 
0.70 
0.50 

16 
14 
12 
10 
8 
6 
4 
2 

0 
0 

*>•* 

-^, 

/ 

/ 

^•sv 

y 

/ 

>sv 

/ 

X 

/ 

1     0.2  0.3  0.5      1.0       23      5710      2JC 

Commercial  Specification  References. 

Joint  Army-Navy  Specification  JAN-C-5,  Capacitors, 

Mica-Dielectric,  Fixed. 
RMA  Standards  Proposal  158A. 


Frequency  to  megacycles 

FIG.  12.    Current-carrying  Capacity  for  10  Deg 

Cent    Temperature    Rise   (Courtesy   of   Solar 

Mfg.  Corp.) 


21.  CERAMIC  DIELECTRIC  CAPACITORS 

Ceramic  capacitors  are  neither  new  nor  the  result  of  any  accidental  discovery.  They 
are  the  direct  result  of  long  research  in  the  early  1900's  when  German  scientists  noted  the 
unusual  characteristics  of  titanate  ceramic  materials. 

Europe  has  never  had  a  domestic  source  of  mica  and  was,  therefore,  faced  with  a  serious 
shortage  before  World  War  I.  This  shortage  focused  attention  and  accelerated  research 


3-68 


RESISTORS;   INDUCTORS;   CAPACITORS 


into  the  possibilities  of  developing  a  suitable  substitute.  This  problem  in  ceramic  research 
took  German  scientists  many  years  to  solve.  In  the  early  1930's,  a  ceramic  titanate  mate- 
rial was  finally  developed  that  was  controllable  in  production  quantities  and,  at  the  same 
time,  would  retain  stable  characteristics.  These  ceramic  dielectric  units  were  quickly  used 
in  substantial  quantities  by  European  radio  and  electronic  industries. 

The  ceramic  dielectric  because  of  its  negative  temperature  coefficient  was  used  in  tem- 
perature-compensating capacitors  in  oscillator  circuits.  As  more  experience  was  gained 
and  the  basic  characteristics  became  more  clearly  understood,  other  forms  of  ceramic 
capacitors  were  used,  including  higher  capacitances,  such  as  by-pass  and  coupling  types 
for  transmitting  and  other  high-voltage  and  high-current  capacitors. 

The  electronics  industry  in  the  United  States  was  slow  to  recognize  the  possibilities  of 
ceramic  capacitors  and  to  utilize  the  general  existing  knowledge  of  European  ceramic 
dielectric  materials  because  of  the  abundant  supplies  of  mica  which  were  readily  available. 
A  second  retarding  factor  was  the  general  feeling  that  ceramic  dielectrics  in  the  titanate 
group  would  be  far  too  costly  for  our  mass-production  methods. 

Centralab,  a  division  of  Globe  Union,  Inc.,  in  the  early  1930's,  initiated  a  research 
program  to  investigate  the  availability  of  raw  materials  and  the  possibility  of  producing 
ceramic  capacitors  similar  to  the  European  types.  Abundant  domestic  supplies  of  raw 
materials  were  located  which  exhibited  characteristics  superior  to  those  of  the  European 
materials. 

The  first  group  of  capacitors  was  offered  in  capacitance  value  up  to  1000  jujuf  and  in 
controlled  temperature  coefficient  varying  from  a  positive  change  of  100  ppm  to  a  negative 
change  of  750  ppm.  The  dielectric  constant  of  these  materials  varied  from  50  in  the  zero 
temperature  coefficient  group  to  a  maximum  of  95  in  the  group  having  a  negative  tempera- 
ture coefficient  of  750  ppm. 

The  use  of  ceramic  capacitors  was  greatly  accelerated  during  World  War  II  because  of 
the  increased  demand  for  substitutes  for  mica  capacitors,  which  were  in  short  supply 
owing  to  a  shortage  of  high-quality  mica. 

The  fact  that  ceramic  capacitors  possess  low  losses  at  ultra-high  frequencies  makes  them 
ideally  suited  for  application  where  other  types  of  dielectrics  are  not  satisfactory.  They 
are  available  in  both  tubular  and  disk  constructions.  The  disk  constructions  employ  a 

"feed-through"     terminal     arrange- 

M-argto  Silver  electrodes          ment  which  re(juces  lead  inductance 

to  the  absolute  minimum. 

A  typical  construction  for  the  tu- 
bular ceramic  capacitors  is  shown  in 
Fig.  13.  In  this  construction,  the  ca- 
pacitance is  controlled  by  the  dielec- 
tric constant  of  the  ceramic,  by  the 
length,  diameter,  and  wall  thickness 
of  the  ceramic  tube,  and  by  the  sil- 
vered area  of  the  electrodes.  Stabil- 


End 


Ceramic 

dialectric  to  electrode 

FIG.  13.     Construction  Detail:  Ceramic  Capacitor 


Steatite 
tube 


soldered 


ity  of  capacitance  with  temperature  and  applied  voltage  is  obtained  with  the  use  of  low-K 
ceramics  in  the  range  of  50  to  500.  The  dielectric  constant  may  be  increased  through  the  use 
of  titanates,  but  this  is  accompanied  by  a  reduction  in  capacitance  stability  and  a  marked 
increase  in  voltage  coefficient  which  imposes  limitations  on  the  use  of  the  high-.K'  bodies. 

Commercial  Specification  References. 

Joint   Army-Navy   Specification   JAN-C-20,    Capacitors,    Ceramic-Dielectric,   Fixed    (Temperature- 
Compensating)  . 
RMA  Standards  Proposal  157,  Ceramic  Dielectric  Capacitors. 

22.  ELECTROLYTIC  CAPACITORS 

Electrolytic  capacitors  employ  solid  dielectric  media  on  which  an  oxide  film,  produced 
electrochemically,  is  the  dielectric  in  the  presence  of  a  d-c  polarizing  voltage  and  an  ionic 
conducting  medium. 

Investigations  have  shown  that  the  oxides  of  tantalum  and  aluminum  exhibit  desirable 
characteristics.  Tantalum  possesses  an  oxide  with  high  dielectric  constant  and  low 
leakage,  but  economic  factors  have  limited  its  use.  Its  principal  limitations  are  voltage 
rating  and  mechanical  construction  resulting  from  the  use  of  sulfuric  acid  as  the  electrolyte. 
Aluminum  is  the  metal  used  in  all  other  electrolytic  capacitors  because  of  its  low  price 
and  excellent  film-forming  characteristics. 

An  oxide  film  can  be  formed  on  aluminum  by  electrolytic  means  by  immersing  a  ribbon 
of  alnTnirmm  foil  in  an  aqueous  solution  of  boric  acid  and  sodium  borate  and  passing  an 


ELECTROLYTIC  CAPACITORS  3-69 

electric  current  through,  the  solution  with  the  aluminum  forming  the  positive  pole  or  anode. 
Electrolysis  of  the  solution  causes  oxygen  to  be  generated  at  the  positive  pole,  oxidizing 
the  surface  of  the  aluminum.  The  film  thickness  is  a  function  of  the  d-c  formation  voltage. 

The  extremely  thin  oxide  film  formed  on  the  aluminum  anode  offers  a  very  high  resist- 
ance to  further  passage  of  current  if  the  applied  voltage  is  not  increased  above  the  film 
formation  voltage.  A  cell  of  this  nature  inserted  in  a  container  containing  an  aqueous 
electrolyte  takes  the  form  of  the  so-called  wet  electrolytic  capacitor.  The  aluminum  oxide 
film  acts  as  the  dielectric,  the  electrolyte  as  the  cathode,  and  the  container  as  the  contact 
medium  for  the  cathode. 

The  electrolytic  capacitor  has  a  very  high  capacitance  per  unit  volume  as  compared  to 
other  types  of  capacitors,  such  as  the  impregnated-paper  or  mica  dielectric  types. 

The  primary  reason  why  an  electrolytic  capacitor  gives  a  high  capacitance  per  unit 
volume  is  the  extreme  thinness  of  the  dielectric  or  oxide  film.  The  thickness  of  the  oxide 
film  covering  the  aluminum  electrode  is  approximately  2  X  10"5  in.,  and  the  dielectric 
constant  K  of  the  oxide  layer  produced  is  high  (approximately  10)  as  compared  with  2.5 
for  a  mineral  oil  impregnated  paper  capacitor. 

There  are  two  types  of  electrolytic  capacitors,  depending  upon  the  physical  character- 
istics of  the  electrolytes:  the  wet  type,  which  uses  an  aqueous  electrolyte;  and  the  dry  type, 
which  uses  a  viscous  or  paste  electrolyte. 

ANODE  FOIL  TREATMENTS.  The  anode  foil  employed  in  electrolytic  capacitors 
may  assume  several  forms,  depending  upon  design  considerations: 

1.  Plain  foil,  where  the  oxide  film  is  electrochemically  formed  on  the  surface  of  the 
aluminum  foil  without  any  previous  treatment  of  the  foil  surface. 

2.  Etched  foil,  where  the  surface  of  the  aluminum  foil  is  first  treated  chemically  or  eiectro- 
chemically  to  erode  the  surface,  thereby  increasing  the  superficial  area  prior  to  the  film- 
forming  procedures. 

3.  Sprayed  gauze,  where  an  inert  carrier  such  as  chemically  pure  cotton  gauze  is  mechan- 
ically coated  with  aluminum  by  means  of  metal  spraying. 

The  etched  foil  electrode  is  the  most  common  one  in  both  wet  and  dry  types  of  electro- 
lytic capacitors,  although  the  sprayed  gauze  construction  is  becoming  more  common  in 
the  dry  electrolytic  capacitor  types. 

The  hydrochloric  acid  etched  anode  foil  construction  is  used  in  place  of  plain  foil  because 
it  gives  effective  surface  areas  much  greater  than  those  obtained  from  plain  foil  and 
thereby  cuts  the  physical  size. 

CATHODE  OR  COLLECTOR  FOIL  TREATMENTS.    Filming  of  the  cathode  foil  is 
a  very  interesting  phenomenon  which  has  been  observed  in  very  high  gain  foils  in  circuits 
of  high  ripple  currents  and  which  must  be  care- 
fully considered  in  the  use  of  electrolytic  capaci-  /T> • s|  TL 

tors.     The  mechanism  that  causes  cathode  film  \L7    -L    f  j_oad 

formation  can  be  explained  by  the  fact  that  the  i      I 

cathode  foil  under  high  ripple  current  conditions 
is  subject  to  a  reversal  of  ripple  current  due  to  the 

charging  of  the  capacitor  on  the  conducting  part      i  -A      /   -  \ -y     r ^j/    \    IAE. 

of  the  cycle  and  the  discharging  of  the  capacitor     Eia"cEd.c/  j       \         T      V 

into  the  load  on  the  non-conducting  part  of  the      *   I    /         | /         \ (          \ , 

cycle  even  though  the  cathode  never  becomes  ER=peak  ripple  voltage 

positive  with  respect  to  the  anode  polarizing  Ea-c—chargmg  voltage 

potential,   as  illustrated  by  Fig.    14.     As   can  Ed-c Average  voitqge 

be  seen  from  the  diagram,  on  one  part  of  the  IL~  Load  CUPimt 

cycle  the  capacitor  is  charging  and  during  the    FIG.  14.    Mechanism  of  Cathode  Formation 

other   half-cycle   the   capacitor   is   discharging 

through  the  load.    The  result  is  that  the  cathode  becomes  anodized  to  the  value  of  the 

ripple  voltage  impressed  across  the  capacitor.    The  value  of  this  potential  is  a  function 

of  the  power-supply  regulation,  the  capacitor  impedance,  and  the  magnitude  of  the  load 

current.    When  the  capacitance  of  the  cathode  is  reduced  by  formation,  it  produces  in 

effect  a  low  instead  of  a  high  capacitance  in  series  with  the  anode  capacitance  and  reduces 

the  capacitance  of  the  capacitor,  depending  upon  the  degree  of  cathode  film  formation. 

Etched  Cathodes.  Most  commercial  designs  employ  etched  cathode  foil  for  low-voltage 
sections  (less  than  25  working  volts  d-c)  and  for  other  ratings  where  ripple  current  is  high. 
Etched  cathode  foil  is  used  for  the  following  reasons: 

1.  An  etched  cathode  foil  increases  the  effective  cathode  surface  area,  which  reduces 
the  ripple  current  density  to  such  a  value  that  the  cathode  film  does  not  build  up  to  a 
value  exceeding  the  initial  thickness  of  the  cathode  film. 

2.  The  increased  cathode  surface  will  afford  a  much  higher  cathode  capacitance,  and, 
the  higher  the  value  of  the  cathode  capacitance,  the  smaller  will  be  the  reduction  in  initial 


3-70 


RESISTORS,   INDUCTORS,   CAPACITORS 


anode  capacitance  because  of  the  series  connection  of  the  cathode  and  anode  capacitance. 
For  example,  a  100-yuf  15  working  volt  d-c  electrolytic  capacitor  with  a  plain  foil  cathode 
has  a  cathode  capacitance  of  approximately  120  /zf ,  which  would  cut  the  total  capacity  of 
the  unit  to  54.5  juf.  However,  if  an  etched  cathode  foil  was  used,  the  cathode  capacitance 
would  be  approximately  1200  jwf ,  and  the  resultant  total  capacitance  of  the  capacitor  would 
be  93  fd . 

Table  8  gives  the  maximum  root  mean  square  ripple  currents  recommended  in  RMA 
Standards  Proposal  160 A  for  various  capacitance  and  voltage  ratings. 

Table  8.     Maximum  rms  Ripple  Current  in  Milliamperes  at  120  Cycles 


Micro- 
farads 

15v 

25  v 

50  v 

150v 

250  v 

300  v 

350  v 

400  v 

450  v 

10 

80 

90 

145 

150 

150 

20 

90 

160 

180 

180 

180 

180 

30 

135 

180 

200 

200 

200 

200 

40 

190 

190 

200 

200 

200 

200 

50 

200 

200 

200 

200 

200 

200 

60 

135 

200 

200 

200 

200 

200 

200 

70 

150 

200 

200 

200 

200 

200 

200 

80 

200 

200 

200 

200 

200 

200 

200 

90 

200 

200 

200 

200 

200 

200 

100 

200 

200 

200 

200 

200 

200 

200 

300 

300 

300 

300 

300 

300 

400 

400 

400 

400 

500 

500 

500 

500 

600 

625 

600 

550 

650 

700 

700 

580 

700 

750 

800 

625 

750 

900 

900 

700 

850 

1000 

700 

850 

1500 

850 

1000 

2000 

1000 

Table  9.     Commercial  Capaci- 
tance Tolerances 


It  is  very  important  that  electrolytic  capacitors  should  not  be  subjected  to  ripple  cur- 
rents in  excess  of  the  values  listed  in  Table  8 ;  otherwise,  the  capacitors  will  overheat  and 
the  life  will  be  appreciably  reduced. 

CHARACTERISTICS  OF  ELECTROLYTIC  CAPACITORS.  A  knowledge  of  the  basic 
characteristics  of  electrolytic  capacitors  is  essential  for 
^^  C0rrect  use.  These  are: 

Capacitance.  The  capacitance  of  a  dry  electrolytic 
capacitor  is  determined  by  the  surface  area  of  the  anode 
and  the  dielectric  thickness  by  the  formation  voltage. 
With  etched  foil  construction,  the  increase  in  superficial 
area  or  gain  varies  inversely  with  the  formation  voltage. 
There  is  a  problem  of  capacitance  control  with  formation 
voltage,  since  for  lower  voltages  capacitance  per  unit  area 
varies  considerably.  See  Table  9. 
Leakage  Current.  The  leakage-current  characteristic  of  a  dry  electrolytic  capacitor 
represents  the  amount  of  direct  current  flowing  through  the  capacitor  with  its  rated  polariz- 
ing voltage  applied,  and  does  not  include  the  momentary  charging  current.  This  leakage 
current  is  an  indication  of  the  quality  of  the  anode  film.  Commercial  specifications  for 
permissible  leakage  current  may  be  determined  by  the 

formula:  Table  10.    Leakage-current 

I  -  KC  +  0.3  Constant 


D-c  Working 
Voltage 

Capacitance 
Tolerance 

0-  50 
51-350 
351-450 

-10;  +250% 
-10;  +100% 
-10;  +50% 

where  I  is  the  d-c  leakage  in  milliamperes,  K  is  the  constant 
as  shown  in  Table  10,  and  C  is  the  rated  capacitance  in  micro- 
farads. The  leakage  current  is  determined  after  application 
of  rated  d-c  working  voltage. 

D-c  "Working  Voltage.  The  d-c  working  voltage  is  the 
maximum  d-c  voltage  the  capacitor  will  stand  under  con- 
tinuous operation  within  its  normal  temperature  range. 

Peak  Working  Voltage.  The  peak  working  voltage  represents  the  d-c  voltage  plus  the 
peak  a-c  ripple  voltage;  it  refers  to  a  continuous  operating  condition  and  should  not  be 
confused  with  surge  voltage. 


0-c  Rated 
Voltage 

K 

3  to  100 
101  to  250 
251  to  350 
351  to  450 

0.01 
0.02 
0.025 
0.04 

ELECTKOLYTIC  CAPACITOES 


3-71 


Table    11.      Single-voltage 
Ratings. 


D-c  Rated 
Voltage 

D-c  Surge 
Voltage 

3 

4 

10 

12 

15 

20 

25 

40 

50 

75 

150 

185 

250 

300 

300 

350 

350 

400 

400 

450 

450 

500 

Table  12.     P  Factors 


Surge  Voltage.  The  surge  voltage  is  a  short-time  d-c  voltage  rating  that  exceeds  the 
peak  working  voltage  and  approaches  the  film  formation  voltage.  This  voltage  is  limited 
by  the  internal  heating  of  the  capacitor  caused  by  the  rapid  increase  in  leakage  current 
as  shown  by  Fig.  17.  Usage  has  established  surge-voltage  ratings  for  various  d-c  working 
voltages  which  are  listed  in  Table  11  for  a  1000-ohm  circuit 
regulation  resistance. 

Power  Factor.  For  all  practical  purposes,  the  power  factor 
of  an  electrolytic  capacitor  is  the  ratio  between  equivalent 
series  resistance  and  the  capacitance  reactance  at  a  given 
frequency.  It  is  expressed  in  percentage  and  indicates  the 
energy  consumed  by  the  capacitor. 

Equivalent  Series  Resistance.  Equivalent  series  resistance 
is  a  more  useful  characteristic  in  mathematical  equations  re- 
lating to  electrolytic  capacitors.  The  equivalent  series  resist- 
ance represents  the  total  losses.  rs  =  watts  A"2,  where  ra  — 
equivalent  series  resistance  and  i  =  leakage  current.  The 
total  losses  in  an  electrolytic  capacitor  consist  of:  (1)  dielec- 
tric loss  of  oxide  film,  (2)  electrolyte  resistance,  and  (3)  con- 
tact resistance.  The  combined  effect  of  these  losses  is  ex- 
pressed as  the  equivalent  series  resistance  value  necessary  to 
produce  an  i2r  loss  of  the  same  magnitude. 

A  convenient  figure  of  merit  for  evaluating  losses  in  electrolytic  capacitors  is  the  P 
factor,  which  is  expressed  as  the  product  of  the  rated  capacitance  in  microfarads  and 
equivalent  series  resistance  in  ohms,  as  measured  on  a  polarized  capacitance  bridge  at  a 
frequency  of  120  cycles  per  second.  Commercial  capacitors  have  P  factors  which  do  not 
exceed  the  values  shown  in  Table  12  for  the  several  standard  d-c  voltage  ratings.  When 
the  high-  and  low-voltage  sections  are  combined  into  a  single  capacitor  winding,  the  elec- 
trolyte for  the  high-voltage  section  determines  the  P  factor  for  the  low-voltage  section  and 
raises  the  P  factor  over  what  would  be  obtained  with  a  low- voltage  electrolyte.  This  makes 
it  necessary  to  double  the  P  factor  (Table  12)  for  the  low- volt  age  section  in  combination 
with  sections  having  d-c  ratings  exceeding  150  volts. 

Temperature  Effects.  Capacitance,  series  resistance, 
power  factor,  and  impedance  of  electrolytic  capacitors  are  all 
somewhat  affected  by  temperature.  Figure  15  shows  capaci- 
tance and  power  factor  vs.  temperature  for  typical  commer- 
cial electrolytic  capacitors  when  operated  over  a  temperature 
range  of  —20  to  85  deg  cent. 

Aircraft  and  other  special  industrial  applications  require 
dry  electrolytic  capacitors  which  will  maintain  constancy  of 
characteristics  for  temperatures  as  low  as  —40  deg  cent. 
Special  electrolytes  are  available  which  meet  these  require- 
ments; see  Fig.  16. 

Dry  electrolytic  capacitors  are  not  recommended  for  use 
in  ambient  temperatures  exceeding  85  deg  cent  because  of 
rapid  drying  out  of  the  electrolyte. 
Leakage  current  increases  with  temperature,  as  shown  by  curve  Fig.  17  for  a  high-voltage 
filter  capacitor  and  low-voltage  by-pass  section  combined  in  a  single  winding. 

Dry  electrolytic  capacitors  intended  for  operation  at  ambient  temperatures  of  85  deg 
cent  require  the  electrolyte  to  be  heat-stabilized  to  prevent  the  leakage  current  from 
increasing  to  the  point  where  the  capacitor  overheats  and  fails.  The  dotted  curve  of  "Fig. 
18  shows  the  reduction  in  leakage  current  at  rated  d-c  voltage  for  various  temperatures 
in  comparison  with  the  leakage  current  for  these  same  temperatures  before  heat  sta- 
bilizing. 

Early  failure  of  dry  electrolytic  capacitors  in  electronic  equipment  is  often  due  to  failure 
to  allow  for  excessive  temperature  in  the  electrolytic  capacitor  specifications. 

R-f  Impedance.  Multiple-section  concentrically  wound  dry  electrolytic  capacitors 
employ  a  common  cathode  which  gives  rise  to  coupling  in  the  cathode  circuit  because  of 
'  the  common  current  paths  as  shown  by  Fig.  19.  In  circuits  where  common  coupling 
causes  circuit  unstability,  a  swedged  cathode  is  employed.  This  construction  effectively 
cuts  the  common  r-f  impedance  to  a  negligible  value  by  the  extension  of  the  cathode  foil 
with  the  turns  swedged  together,  which,  in  effect,  gives  a  non-inductive  cathode  construc- 
tion since  each  cathode  or  ground  terminal  is  directly  under  its  corresponding  anode  ter- 
minal. This  construction  is  similar  as  far  as  the  cathode  is  concerned  to  the  non-inductive 
winding  shown  in  Fig.  6  for  impregnated-paper  capacitors. 


Rated  Voltage 

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3000 

10 

1500 

15 

1200 

25 

500 

50 

400 

150 

300 

250 

250 

300 

250 

350 

250 

400 

250 

450 

250 

3-72 


RESISTORS,  INDUCTORS,   CAPACITORS 


Gas  Pressure, 
expansion.    This 

100 
90 
80 
70 
£60 

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30 
20 
10 

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Electrolytic  capacitors  should  be  supplied  with  a  libei 
extra  space  is  to  accommodate  any  sudden  generation  o 

al  space  for  gas 
f  gas  which  may 

50 
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40 
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"'T.mp.tSu™  in  f,,,f=S4l§Sr,d°.  '°                   T«np.r.tu«  In  d.l»M  cuflgnKl, 

FIG.  15.     Capacitance  and  %  Power  Factor  vs  Temperature  of  Electrolytic  Capacitor  (Courtesy  Solar 

Mfg.  Corp.) 

be  liberated  as  the  result  of  improper  uses  of  the  capacitor.  In  addition,  most  electrolytics 
are  supplied  with  built-in  vents  which  prevent  capacitors  from  exploding  when  they  are 
improperly  used,  as  being  accidentally  connected  across  alternating  current. 


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80 


FIG.  16.     Variation  of  Characteristics  of  Low-temperature  Electrolytic  Cap) 
(16  /if— 350  WVDC— Courtesy  of  Solar  Mfg.  Corp.) 


100    g. 
<5 
[tor  with  Temperature 


Polarity.  Polarized,  types  of  dry  electrolytic  capacitors  are  designed  for  use  in  d-c  or 
intermittent  d-e  circuits  produced  by  rectifying  alternating  current.  D-c  polarized  capaci- 
tors should  not  be  subjected  to  reversed  polarity,  as  the  heavy  current  passing  through  the 
capacitor  under  this  condition  will  raise  its  internal  temperature  and  seriously  damage  it. 


10 


20       30       40        50       60       70 

Temperature  in  degrees  centigrade 


90 


FIG.  17.    D-c  Leakage  vs  Temperature  —  Dry  Electrolytic  Capacitors 


BIBLIOGEAPHY 


3-73 


Non-polarized  Types.  This  term  applies  to  dry  electrolytic  capacitors  constructed  with 
two  formed  electrodes  so  that  they  function  equally  well  with  direct  current  impressed 
on  either  electrode  irrespective  of  polarity.  This  does  not  mean  that  they  can  be  used  on 
alternating  current  continuously.  Non-polarized  capacitors  are  for  applications  where 
the  d-c  voltage  supply  might  become  re- 
versed and  remain  so  indefinitely.  A  non- 
polarized capacitor  is  equivalent  to  two 
polarized  capacitors  connected  in  series 
opposition. 

A-c  Motor-starting  Capacitors.  Non- 
polarized dry  electrolytic  capacitors  may 
be  used  for  intermittent  duty  in  a-c  cir- 
cuits such  as  for  motor  starting,  and  they 
are  known  as  motor-starting  capacitors. 
They  are  ideal  for  intermittent  duty  when 
used  within  the  manufacturers'  limits  for 
operating  voltage,  temperature,  and  duty 
cycle. 

Since  the  duty  cycle  is  based  on  internal 


Leakage  current  fa  mJTItamps 
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20 


100 


30       40    .    50        60       70       80        90 
Temperature  in  degrees  centigrade 

FIG.  18.     Effect  of  Heat  Stabilization  of  Electrolyte 
(30  Mf— 450  WVDC— Courtesy  of  Solar  Mfg.  Corp.) 


heating  of  the  capacitor,  it  is  possible  to  vary  the  number  of  application  periods  of  voltage 
with  the  period  of  duration  of  voltage  so  that  the  product  is  a  constant.  The  manu- 
facturers' guarantee  is  usually  twenty  0.5-sec  starts  per  hour. 

The  normal  maximum  operating  temperature  of  these  capacitors  is  65  deg  cent.    They 
may  be  successfully  operated  up  to  85  deg  cent  provided  that  the  duty  cycle  and  maximum 

voltage   conditions   are   adjusted   ac- 

tt  ©    t  ^ 

Anode       T    T         Anode          ?  Anode^) 

l"NiU>'MUHiNt'UiUNtUni'HHmnn\ © 


Start  of 
winding 


r. 


.  Coupling  path 


(i)  Anode  tab  or  riser 

©Cathode  tab 

©  Anode  foils,  three 

(4)  Contact  foil  or  misscalled  cathode  foil 

©  Cathode,  electrolyte-impregnated  paper 

FIG.  19.     Schematic  Diagram  of  Three  Section  Dry 
Electrolytic  Capacitor  Winding 


cordingly.  However,  it  must  be  noted 
that  operating  at  higher  than  normal 
temperatures  decreases  the  life  consid- 
erably as  it  dries  out  the  electrolyte  at 
an  accelerated  rate.  Operation  of  ca- 
pacitors at  low  temperatures  cannot 
harm  capacitors  because  any  change 
in  characteristics  is  only  temporary  at 
subzero  temperatures.  The  increase 
in  power  factor  represents  an  increased 
resistance  loss  when  the  capacitor  is  in 
operation,  and  this  creates  sufficient 
heat  to  warm  up  the  capacitor  and 
quickly  return  it  to  normal  operating 
conditions. 


Commercial  Specification  Reference.  Where  more  specific  information  and  data  are 
required  for  testing  methods  and  procedures  and  for  specific  capacitor  ratings,  the  following 
are  suggested: 

Joint  Army-Navy  Specification  JAN-C-62,  Capacitors,  Dry  Electrolytic,  Polarized. 
EMA  Standards  Proposal  160,  Polarised  Dry  Electrolytic  Capacitors. 


BIBLIOGRAPHY 

A.I.E.E.  Standards  for  Capacitors,  Approved  Standard  18,  June  1934. 

Radio  Instruments  and  Measurements,  Circular  of  the  Bureau  of  Standards  74,  Government  Printing 

Office,  1924. 
Maloff,    I.   G.,   Mica  Condensers  in  High-frequency  Circuits,  Proc.  I.R.E.,   Vol.   20,   647    (April 

1932). 
Kouenhoven,  W.  B.,  and  Lemmon,  C.  L.,  Phase  Defect  Angle  of  an  Air  Capacitor,  /.  A.I.E.E.,  Vol.  49, 

No.  11,  945  (November  1930). 
Field,  R.  F.,  An  Equal-arm  Capacitance  Bridge,  General  Radio  Experimenter,  Vol.  4,  No.  8,  1  (January 

1930). 
Burke,  C.  T.,  Substitution  Method  for  the  Determination  of  Resistance  of  Inductors  and  Capacitors 

at  Radio  Frequencies,  Trans.  A.I.E.E.,  Vol.  46,  483  (May  1927). 
Gemant,  A.,  Liquid  Dielectrics,  Jonn  Wiley,  1933. 
Morgan,  S.  0.,  and  White,  A.  H.,  The  Dielectric  Constant  and  Power  Factor  of  Rosin  Oil  and  Ethyl 

Abietate,  J.  Franklin  Inst.,  Vol.  213,  No.  3,  313  (March  1932). 

Hoch,  E.,  Power  Losses  in  Insulating  Materials,  Bett  Sys.  Tech.  J.,  Vol.  1,  No.  2,  110  (November  1922). 
Benedict,  R.  R.,  Behavior  of  Dielectrics,  Trans.  A.I.E.E.,  Vol.  49,  739  (April  1930). 
Lewis,  A   B.,  Hall,  E.  L.,  and  Caldwell,  F.  R.,  Some  Electrical  Properties  of  Foreign  and  Domestic 

Micas,  Bur.  Stand.  J.  Research,  Vol.  7,  No.  2,  403  (August  1931). 
Siegmund,  H.  0.,  The  Aluminum  Electrolytic  Condenser,  Bell  Laboratories  Reprint  349,  Trans.  Electro- 

chem.  Soc.,  Vol.  53,  203  (1928). 
Godsey,  F.  W.,  Jr.,  Film  Characteristics  of  Electrolytic  Condensers,  Trans.  A.I.E.E.,  Vol.  51,  432 

(Jane  1932). 


3-74  RESISTORS,  INDUCTORS,    CAPACITORS 

Godsey,  P.  W.,  Jr.,  A-C  Capacity  of  Electrolytic  Condensers,  Trans.  Electrochem.  Soc.,  Vol.  61,  515 

(April  1932). 
Godsey,  P.  W.,  Jr.,  Cathodic  Films  in  Electrolytic  Condensers,  Trans.  Electrochem.  Soc.,  Vol.  63,  223 

(1933). 

Godsey,  P.  W.,  Jr.,  Potential  Gradients  in  Anodic  Films,  Trans.  Electrochem.  Soc.,  Vol.  61,  549  (1932). 
Lilienfeld,  Applet  on,  Smith,  and  Nieh,  Studies  of  Fully  Organized  Anodic  Layers  on  Aluminum 

Trans.  Electrochem.  Soc.,  Vol.  61,  531  (1932). 

Godsey,  P.  W.,  Jr.,  Power  Losses  in  Electrolytic  Condensers,  Trans.  A.I.E.E.,  Vol.  51,  439  (June  1932). 
Christopher,  A.  J.,  and  Kater,  J.  A.,  Mica  Capacitors  for  Carrier  Telephone  Systems,  Trans.  A.I.E  E 

Vol.  65,  670  (October  1946). 
Murphy,  E.  J.,  and  Morgan,  S.  O.,  The  Dielectric  Properties  of  Insulating  Materials,  Bell  Sys.  Tech.  J.t 

Vol.  16,  493  (October  1937). 
MacLeod,  H.  J.,  The  Variation  with  Frequency  of  the  Power  Loss  in  Dielectrics,  Phys.  Rev.  (2),  Vol.  21 

(January  1923). 

Brotherton,  M.,  Paper  Capacitors  under  Direct  Voltage,  Proc.  I.R.E.,  Vol.  32,  139  (March  1944). 
McLean,  D.  A.,  Edgerton,  L.,  Kohman,  G.  T.,  and  Brotherton,  M.,  Paper  Dielectrics  Containing 

Chlorinated  Impregnants,  Industrial  and  Eng.  Chem.,  Vol.  34,  101  (January  1942). 
Palmer,  H.  B.,  Capacitance  of  a  Parallel  Plate  Capacitor  by  the  Schwartz-Christoffel  Transformation 

Trans.  AJ.E.E.,  Vol.  56,  363  (March  1937). 
Reed,  M.,  Effect  of  Stray  Capacities  to  Ground  in  Substitution  Measurements,  diagrams,  Wireless 

Eng.,  Vol.  13,  248  (May  1936). 
Field,  R.  F.,  and  Sinclair,  D.  B.,  Method  for  Determining  the  Residual  Inductance  and  Resistance  of 

a  Variable  Air  Condenser  at  Radio  Frequencies,  bibliography,  diagrams,  Proc.  I.R.E.,  Vol.  21,  255 

(February  1936) . 

McDonald,  L.  J.,  Contours  of  Capacitor  Rotor  Plates,  Electronics,  Vol.  18,  126  (March  1945). 
Boella,  M.,  Direct  Measurement  of  the  Loss  Conductance  of  Condensers  at  High  Frequencies,  Proc. 

I.R.E.,  Vol.  20,  421  (April  1938). 

Direct-reading  Condenser  for  Substitution  Measurements,  Gen.  Radio  Exp.,  Vol.  10  (March  1936). 
Michaelson,  H.  B.,  Gas-filled  and  Vacuum  Capacitors,  illustrations,  Electronics,  Vol.  17, 124  (September 

1944). 

Sinclair,  D.  B.,  High-frequency  Model  of  Precision  Condenser,  Gen.  Radio  Exp.,  Vol.  12  (October- 
November  1939). 
Griffiths,  W.  H.  P.,  Law  Linearity  of  Semicircular  Plate  Variable  Condensers,  diagrams,  Wireless  Eng., 

Vol.  22,  107  (March  1945). 
Field,  R.  F.,  and  Sinclair,  D.  B.,  Method  for  Determining  the  Residual  Inductance  and  Resistance  of 

a  Variable  Air  Condenser  at  Radio  Frequencies,  bibliography,  diagrams,  Proc,  I.R.E,,  Vol.  24,  255 

(February  1936). 
Green,  A.  P.,  and  McComb,  C.  T.,  Resonance  in  Mica  Capacitors,  Electronics,  Vol.  17,  119  (March 

1944). 
Brinkmann,  C.,  Self -discharge  and  Time  Constant  of  the  High-voltage  Oiled-paper  Condenser,  abstract, 

Wireless  Eng.,  Vol.  20,  449  (September  1943). 
Schick,  W.y  Temperature  Coefficient  of  Capacitance;  Its  Measurement  in  Small  Radio  Condensers, 

bibliography,  illustrations,  diagrams,   Wireless  Eng.,  Vol.  21,  65   (April  1944) ;  Discussion,  T.  J. 

Rehfisch,  Vol.  21,  175  (February  1944). 

Coursey,  P.  R.,  Thermal  Stability  of  Condensers;  Ceramic  Dielectrics  and  Their  Use  at  Low  Tempera- 
tures, Wireless  Eng.,  Vol.  15,  247  (May  1938). 

Floyd,  G.  H.,  Vacuum  Capacitors,  illustrations,  diagrams,  Proc.  I.R.E. ,  Vol.  32,  463  (August  1944). 
Proctor,  R.  F.,  Variable  Air  Condensers;  Determination  of  Their  Residual  Parameters,  diagrams, 

Wireless  Eng.,  Vol.  17,  257  (June  1940). 
Attwood,  S.  S.,  and  Bixby,  W.  H.,  Breakdown  and  Time-Lag  of  Dielectric  Materials,  J.  Franklin  Inst., 

March  1943. 

Balsbaugh,  J.  C.,  Assaf,  A.  G.,  and  Oncley,  J.  L.,  Dielectric  Properties  of  Hydrocarbons  and  Hydro- 
carbon Oils,  Industrial  and  Eng.  Chem.,  January  1942,  pp.  92-100. 
Barringer,  L.  E.,  The  Relation  of  Chemical  and  Physical  Structure  to  Dielectric  Behavior,  Trans. 

Electrochem,  Soc.,  Vol.  LXV  (1934). 

Burnett,  J.  H.,  Liquid,  Viscous,  and  Solid  Dielectrics,  Electric  Manufacturing,  August  1943. 
Clark,  F.  M.,  The  Development  and  Application  of  Synthetic  Liquid  Dielectrics,  Trans.  Electrochem. 

Soc.,  Vol.  LXV  (1934). 
Clark,  F.  M.,  and  Raab,  E.  L.,  Electrical  Stability  of  Mineral  Oil-Treated  Dielectrics,  Industrial  and 

Eng.  Chem.,  January  1942,  pp.  110-116. 
McLean,  D.  A.,  and  Egerton,  L.,  Paper  Capacitors  Containing  Chlorinated  Impregnants.    Stabilization 

by  Anthraquinine,  Industrial  and  Eng.  Chem.,  Vol.  37  (January  1945). 
Cornell,  J.  L,  Metallized  Paper  Capacitors,  The  Solar  System,  Vol.  IV,  No.  4  (November-December 

1946),  Solar  Mfg.  Corp. 
Berberich,  L.  J.,  Fields,  C.  V.,  Marbury,  R.  E.,  Characteristics  of  Chlorinated  Impregnants  in  D-c 

Paper  Capacitors,  AJ.E.E.  Technical  Paper  44-165  (May  1944). 
Karplus,  Edward,  Wide  Range  Tuned  Circuits  and  Oscillators  for  High  Frequencies,  Proc.  I.R.E.,  Vol. 

,33,  426  (July  1945). 
Piper,  J.  D.,  Kerstein,  N.  A.,  and  Fleiger,  A.  G.,  Oil  Impregnated  Paper,  Industrial  and  Eng.  Chem., 

September  1937,  pp.  104071043. 

Siegmund,  H.  O.,  The  Aluminum  Electrolytic  Condenser,  Bell  Sys.  Tech.  J.,  January  1929. 
White,  A.  H.,  and  Morgan,  S.  O.,  The  Dielectric  Properties  of  Chlorinated  Diphenyls,  J.  Franklin 

Inst.,  November  1933,  pp.  635-644. 
Vhitehead,  J.  B.,  Liquid  Insulators,  T. 
limmerman,  C.  L,  The  Aluminum  Elec 
Jerberich,  L.  J.,  and  Friedman,  Raymond,  tttaouization  ot  < 

Industrial  and  Eng.  Chem.,  Vol.  40,  117  (January  1948). 
Golding,  E.  W.,  Electrical  Measurements  and  Measuring  Instruments,  Sir  Isaac  Pitman  &  Sons,  Ltd., 

London. 

Georgiev,  Alexander  M.,  The  Electrolytic  Capacitor,  Murray  Hill  Books. 
Brotherton,  M.,  Capacitors — Their  Use  in  Electronic  Circuits,  D.  Van  Nostrand  Co. 
Nersey,  Philip  R.,  Electrolytic  Condensers,  Chapman  Hall,  2nd  Ed.,  London,  1939. 
Deeley,  Paul  McKnight,  Electrolytic  Capacitors,  Cornell-Dubilier,  1938. 

Ooursey,  P.  R.,  Electrical  Condensers,  Their  Construction,  Design  and  Industrial  Uses,  Pitman,  1927. 
.Schwaiger,  A.,  Theory  of  Dielectrics,  John  Wiley,  1932. 
fWhitehead,  J.  B.,  Impregnated  Paper  Insulation,  John  Wiley  and  Sons,  1935. 


SECTION  4 
ELECTRON  TUBES 


THERMIONIC  VACUUM  TUBES 
ART.  BY  A.  P.  KAUZMANN  PAGE 

1.  Principles  of  Operation 02 

2.  Classifications 03 

3.  Definitions 03 

4.  Methods   of   Measuring  Tube   Currents 

and  Parameters   08 

5.  Vacuum-tube  Operation 14 

6.  Typical     Vacuum-tube     Characteristic 

Curves 31 

MAGNETRONS 
BY  W.  B.  HEBENSTREIT 

7.  The  Non-oscillating  Magnetron 40 

8.  The  Oscillating  Magnetron 40 

9.  Operation  of  the  Traveling-wave  Magne- 

tron       42 

KLYSTRONS 
BY  A.  L.  SAMUEL 

10.  Klystrons     (Employing     Transit     Time 

Bunching) 51 

11.  Reflex  Klystrons 54 

12.  Tube  Types 57 


GASEOUS  CONDUCTION  TUBES 

ART.  BY  D.  S.  PECK  PAGE 

13.  Gaseous  Conduction 58 

14.  Thyratron  Tubes 60 

15.  Voltage  Limits  of  Thyratrons 62 

16.  Current  Limits  of  Thyratrons : .  .  63 

17.  Control  Characteristics 65 

18.  Pulse  Thyratrons 69 

19.  Installation     and     Operation     of     Hot- 

cathode  Thyratron  Tubes 71 

20.  Cold-cathode  Tubes 72 

21.  Pool-cathode  Tubes 75 

X-RAY  TUBES 

BY  S.  REID  WARREN,  JR. 

22.  General  Physical  Requirements 81 

23.  Tubes  for  X-ray  Therapy 83 

24.  Tubes  for  Medical  Roentgenography  and 

Roentgenoscopy 86 

25.  Tubes   for   Industrial   Roentgenography 

and  Fluoroscopy,  and  for  X-ray  Dif- 
fraction   89 


4-01 


ELECTRON  TUBES 
THERMIONIC  VACUUM  TUBES 

By  A.  P.  Kauzmann 

As  dealt  with  in  articles  1-6,  vacuum  tubes  in  operation  are  characterized  by  a  source 
of  electron  emission;  the  conduction  through  a  vacuous  space,  which  may  or  may  not 
contain  sufficient  gas  to  affect  the  conduction,  of  a  current  between  the  source  of  emission 
and  one  or  more  other  electrodes  by  means  of  the  emitted  electrons;  and  the  varying 
of  this  current  by  means  of  variations  in  electrode  potential  to  produce  an  electrical 
response  in  an  associated  circuit.  Thus,  phototubes  and  cathode-ray  tubes,  though  strictly 
vacuum  tubes,  are  excluded  (see  Section  15).  Grid-controlled  gas-discharge  tubes  (see 
articles  13-21)  and  mercury-pool-type  rectifiers  (see  article  21)  are  treated  later  in  the 
section  because  their  special  properties  and  applications  require  special  treatment. 

1.  PRINCIPLES  OF  OPERATION 

HIGH-VACtTCTM  TUBES.  The  simplest  form  of  high-vacuum  tube  is  the  diode,  con- 
sisting of  a  thermionic  cathode  and  an  anode.  The  cathode  is  heated  to  a  temperature,  de- 
pending on  its  nature,  at  which  electrons  are  emitted  from  its  surface  into  the  surround- 
ing vacuous  space.  When  the  anode  is  placed  at  a  potential  positive  with  respect  to  the 
cathode  some  of  these  emitted  electrons  are  caused  to  flow  to  the  anode  under  the  in- 
fluence of  the  electrostatic  field,  thus  constituting  a  current  flowing  from  cathode  to 
anode  in  the  external  circuit,  since  the  electrons  are  negatively  charged.  Those  electrons 
which  are  not  drawn  to  the  anode  return  to  the  cathode. 

Because  of  the  mutual  repulsion  between  the  like  charges  of  the  electrons,  only  a  def- 
inite number  of  them  may  be  accommodated  in  the  space  between  cathode  and  anode 
at  any  given  anode  potential,  and  therefore  the  current  flowing  is  definitely  limited. 
This  limiting  effect  is  called  space  charge,  and  the  current  is  said  to  be  space-charge  limited. 
As  the  anode  potential  is  increased  the  current  increases  until  the  total  emission  of  the 
cathode  is  drawn  to  the  anode,  beyond  which  point  the  current  is  said  to  be  temperature 
limited. 

The  relation  between  current  and  potential  under  conditions  of  space-charge  limitation 
may  be  represented  approximately  by  the  equation 


where  K\  is  a  constant  depending  on  the  physical  dimensions  of  the  tube. 

Since  the  electron  flow  to  the  anode  depends  on  the  electrostatic  field  in  the  neighbor- 
hood of  the  cathode,  the  current  may  be  controlled  in  part  by  the  potential  of  another 
electrode  in  a  position  to  influence  this  field.  In  the  simple  triode  this  additional  electrode 
consists  of  a  gric?-like  structure  placed  between  cathode  and  anode.  Because  of  its  prox- 
imity to  the  cathode,  the  grid  has  more  control  of  the  field  near  the  cathode  than  the 
anode  has,  but  because  of  its  open  structure  most  of  the  electrons  pass  through  to  the 
anode.  When  the  grid  is  operated  at  a  negative  potential  with  respect  to  the  cathode, 
none  of  the  electrons  are  taken  by  the  grid. 

The  relation  between  anode  current  and  grid  and  anode  potentials  is  usually  repre- 
sented by  the  equation 

ib  =  K*(eb  +  vec^  (1) 

where  K*  is  a  constant  depending  primarily  on  the  cathode  and  grid  dimensions,  ju  is  the 
amplification  factor  of  the  tube  (determined  by  the  grid  structure  and  grid-anode  spacing)  , 
and  rj  is  usually  between  1.5  and  2.5. 

Other  electrodes,  usually  grids,  may  be  added  to  the  structure,  but  the  fundamental 
principles  involved  remain  the  same.  For  applications  of  high-vacuum  tubes  see  Sections 
7,  16,  17,  19,  20,  and  21. 

GAS-FILLED  TUBES.  In  a  gas-filled  diode,  the  space  contains  sufficient  gas  at  a  low 
pressure  to  cause  an  appreciable  fraction  of  the  electrons  passing  between  cathode  and 

4-02 


DEFINITIONS  4-03 

anode  to  collide  with  the  gas  molecules  and  thereby  ionize  them.  The  dislodged  electrons 
pass  on  to  the  anode  as  additional  anode  current,  while  the  positively  charged  ions  are 
drawn  to  the  cathode,  but  at  a  much  lower  velocity,  owing  to  their  greater  mass,  than 
the  electrons  possess.  Because  of  this  low  velocity  a  given  positive-ion  current  produces 
a  much  higher  space-charge  density  than  the  same  electron  current.  Therefore,  the  net 
space-charge  density  produced  by  the  two  currents  (the  algebraic  sum  of  the  positive 
and  negative  space  charges)  may  approach  zero,  though  the  electron  current  is  by  far 
the  larger. 

Since  the  limitation  of  current  in  a  high-vacuum  diode  is  due  to  the  negative  space 
charge  and  this  space  charge  is  reduced  by  the  positive  ions,  it  follows  that  the  presence 
of  the  gas  increases  the  current  which  may  flow  at  a  given  anode  potential.  Under  normal 
conditions,  the  discharge  is  unstable  at  potentials  much  in  excess  of  the  ionizing  potential 
of  the  gas,  the  current  increasing  until  limited  by  the  cathode  emission.  For  this  reason, 
the  current  is  usually  limited  by  an  external  resistance  in  series  with  the  anode. 

The  relation  between  applied  voltage  and  current  may  be  expressed  by  the  equation 


where  Eb  is  the  practically  constant  "anode  drop"  of  the  tube  and  r  is  the  load  resistance. 

2.  CLASSIFICATIONS 

Vacuum  tubes  in  general  are  classified  according  to  many  different  structural  and 
electrical  characteristics. 

Cathodes  are  directly  heated  if  the  actual  emitter  is  also  the  resistance  element  which 
supplies  the  heat,  and  indirectly  heated  if  the  heat  is  supplied  by  conduction  or  radiation 
from  a  resistance  element.  The  directly  heated  cathode  is  more  efficient  than  the  indirectly 
heated,  but  for  many  applications  may  not  be  heated  by  alternating  current.  The 
cathode  material  may  be  tungsten,  thoriated  tungsten,  or  oxide-coated  metal.  Tungsten  is 
the  least  efficient  and  is  generally  used  only  in  high-voltage  tubes.  Thoriated  tungsten 
is  much  more  efficient  than  tungsten  and  is  used  in  many  medium-voltage  tubes  (500 
to  2000  volts  anode  potential)  and  some  low-voltage  tubes;  at  high  voltages  the  emission 
life  may  be  short.  Neither  of  these  materials  is  used  for  indirectly  heated  cathodes. 
Oxide-coated  cathodes  are  the  most  efficient  but  (except  in  gas-filled  tubes)  are  com- 
monly used  only  in  low-voltage  tubes  because  of  troubles  from  grid  emission. 

Anodes  are  radiation  cooled,  or  air  cooled,  if  no  other  cooling  means  is  provided,  or  water 
cooled  if  provided  with  means  for  circulating  water  about  or  through  the  anode.  Only 
high-  voltage  high-power  (more  than  6000  volts,  2500  watts  rating)  tubes  are  water  cooled, 
except  for  special  high-frequency  tubes. 

Rectifiers  are  high  vacuum  or  gas  filled.  For  high-power  high-voltage  applications,  gas- 
filled  tubes  are  usual,  though  high-vacuum  tubes  are  used  above  20,000  volts.  For  radio 
receivers  both  are  used,  gas-filled  tubes  predominating  in  other  low-  voltage  applications. 

Triodes  are  classified  according  to  voltage  and  power  rating,  amplification  factor,  and 
special  features  such  as  designs  for  high-frequency  oscillators,  freedom  from  mechanical 
disturbance,  etc.  There  are  so  many  triodes  of  varying  characteristics  that  almost  any 
requirement  can  be  met. 

Multigrid  tubes  have  been  developed  largely  for  specialized  radio  purposes,  but  the 
screen-grid  tubes,  including  suppressor-grid  pentodes,  have  wide  application.  In  the  screen- 
grid  tube,  the  grid  is  screened  or  shielded  from  the  anode,  thus  greatly  reducing  the  feed- 
back capacitance  between  plate  and  grid.  For  high-frequency  voltage  amplification, 
these  tubes  are  used  almost  exclusively. 

3.  DEFINITIONS 

Vacuum  Tube.  (Electron  Tube.)  A  vacuum  tube  is  a  device  consisting  of  an  evacuated 
enclosure  containing  a  number  of  electrodes  between  two  or  more  of  which  conduction  of 
electricity  through  the  vacuum  or  contained  gas  may  take  place. 

NOTE:  The  term  is  used  in  a  more  restricted  sense  to  mean  a  device  of  this  nature  designed  for 
such  use  as  amplifier,  rectifier,  modulator,  or  oscillator. 

Gas  Tube.  A  gas  tube  is  a  vacuum  tube  in  which  the  pressure  of  the  contained  gas  or 
vapor  is  such  as  to  affect  substantially  the  electrical  characteristics  of  the  tube. 


4-04  ELECTRON  TUBES 

Mercury-vapor  Tube.  A  mercury-vapor  tube  is  a  gas  tube  in  which  the  active  con- 
tained gas  is  mercury  vapor. 

High -vacuum  Tube.  A  high-vacuum  tube  is  a  vacuum  tube  evacuated  to  such  a  degree 
that  its  electrical  characteristics  are  essentially  unaffected  by  gaseous  ionization. 

Thermionic  Tube.  A  thermionic  tube  is  a  vacuum  tube  in  which  one  of  the  electrodes 
is  heated  for  the  purpose  of  causing  electron  or  ion  emission. 

Phototube.  (Photoelectric  Tube.)  A  phototube  is  a  vacuum  tube  in  which  one  of  the 
electrodes  is  irradiated  for  the  purpose  of  causing  electron  emission 

Cathode-ray  Tube.  A  cathode-ray  tube  is  a  vacuum  tube  in  which  the  electron  stream 
is  directed  along  a  confined  path  to  produce  non-electrical  effects  on  the  object  upon 
which  the  electrons  impinge. 

NOTE:  This  classification  includes  cathode-ray  oscillograph  tubes,  similar  devices  for  television 
reception,  electron  microscopes,  etc. 

Cathode-ray  Oscillograph  Tube.  A  cathode-ray  oscillograph  tube  is  a  cathode-ray 
tube  in  which  the  movement  of  an  electron  beam,  deflected  by  means  of  applied  electric 
and/or  magnetic  fields,  indicates  the  instantaneous  values  of  the  actuating  voltages  and/or 
currents. 

Diode.    A  diode  is  a  two-electrode  vacuum  tube  containing  an  anode  and  a  cathode. 

Triode.  A  triode  is  a  three-electrode  vacuum  tube  containing  an  anode,  a  cathode,  and 
a  control  electrode. 

Tetrode.  A  tetrode  is  a  four-electrode  vacuum  tube  containing  an  anode,  a  cathode,  a 
control  electrode,  and  one  additional  electrode  ordinarily  in  the  nature  of  a  grid. 

Pentode.  A  pentode  is  a  five-electrode  vacuum  tube  containing  an  anode,  a  cathode, 
a  control  electrode,  and  two  additional  electrodes  ordinarily  in  the  nature  of  grids. 

Hexode.  A  hexode  is  a  six-electrode  vacuum  tube  containing  an  anode,  a  cathode,  a 
control  electrode,  and  three  additional  electrodes  ordinarily  in  the  nature  of  grids. 

Heptode.  A  heptode  is  a  seven-electrode  vacuum  tube  containing  an  anode,  a  cathode, 
a  control  electrode,  and  four  additional  electrodes  ordinarily  in  the  nature  of  grids. 

Octode.  An  octode  is  an  eight-electrode  vacuum  tube  containing  an  anode,  a  cathode, 
a  control  electrode,  and  five  additional  electrodes  ordinarily  in  the  nature  of  grids. 

Multiple-unit  Tube.  A  multiple-unit  tube  is  a  vacuum  tube  containing  within  one 
envelope  two  or  more  groups  of  electrodes  associated  with  independent  electron  streams. 

NOTE:  A  multiple-unit  tube  may  be  so  indicated,  as,  for  example;  duodiode,  duotriode,  diode-pentode, 
duodiode-triode,  duodiode-pentode,  and  triode-pentode. 

Cathode.     A  cathode  is  an  electrode  which  is  the  primary  source  of  an  electron  stream. 

Filament.  A  filament  is  a  cathode  of  a  thermionic  tube,  usually  in  the  form  of  a  wire  or 
ribbon,  to  which  heat  may  be  supplied  by  passing  current  through  it. 

Indirectly  Heated  Cathode.  (Equipotential  Cathode,  TJnipotential  Cathode.)  An  in- 
directly heated  cathode  is  a  cathode  of  a  thermionic  tube  to  which  heat  may  be  supplied 
by  an  independent  heater  element. 

Heater.  A  heater  is  an  electric  heating  element  for  supplying  heat  to  an  indirectly 
heated  cathode. 

Control  Electrode.  A  control  electrode  is  an  electrode  on  which  a  voltage  is  impressed 
to  vary  the  current  flowing  between  two  or  more  other  electrodes. 

Grid.  A  grid  is  an  electrode  having  one  or  more  openings  through  which  electrons  or 
ions  may  pass. 

Space-charge  Grid.  A  space-charge  grid  is  a  grid  which  is  placed  adjacent  to  the  cath- 
ode and  positively  biased  so  as  to  reduce  the  limiting  effect  of  space  charge  on  the  current 
through  the  tube. 

Control  Grid.  A  control  grid  is  a  grid,  ordinarily  placed  between  the  cathode  and  the 
anode,  for  use  as  a  control  electrode. 

Screen  Grid.  A  screen  grid  is  a  grid  placed  between  a  control  grid  and  an  anode,  and 
usually  maintained  at  a  fixed  positive  potential,  for  the  purpose  of  reducing  the  electro- 
static influence  of  the  anode  in  the  space  between  the  screen  grid  and  the  cathode. 

Suppressor  Grid.  A  suppressor  grid  is  a  grid  which  is  interposed  between  two  electrodes 
(usually  the  screen  grid  and  plate),  both  positive  with  respect  to  the  cathode,  in  order  to 
prevent  the  passing  of  secondary  electrons  from  one  to  the  other. 

Anade.     An  anode  is  an  electrode  to  which  a  principal  electron  stream  flows. 

Plate.     Plate  is  a  common  name  for  the  principal  anode  in  a  vacuum  tube. 

Electron  Emission.  Electron  emission  is  the  liberation  of  electrons  from  an  electrode 
into  the  surrounding  space.  Quantitatively,  it  is  the  rate  at  which  electrons  are  emitted 
from  an  electrode. 


DEFINITIONS  4-05 

Thermionic  Emission.  Thermionic  emission  is  electron  or  ion  emission  due  directly  to 
the  temperature  of  the  emitter. 

Secondary  Emission.  Secondary  emission  is  electron  emission  due  directly  to  the  im- 
pact of  electrons  or  ions. 

Grid  Emission.     Grid  emission  is  electron  or  ion  emission  from  a  grid. 

Emission  Characteristic.  An  emission  characteristic  is  a  relation,  usually  shown  by  a 
graph,  between  the  emission  and  a  factor  controlling  the  emission  (as  temperature,  voltage, 
or  current  of  the  filament  or  heater). 

Cathode  Current.  Cathode  current  is  the  total  current  passing  to  or  from  the  cathode 
through  the  vacuous  space. 

NOTE:  This  term  should  be  carefully  distinguished  from  heater  current'  and  filament  current. 

Filament  Current.     Filament  current  is  the  current  supplied  to  a  filament  to  heat  it. 

Filament  Voltage.     Filament  voltage  is  the  voltage  between  the  terminals  of  a  filament. 

Heater  Current.     Heater  current  is  the  current  flowing  through  a  heater. 

Heater  Voltage.     Heater  voltage  is  the  voltage  between  the  terminals  of  a  heater. 

Grid  Current.  Grid  current  is  the  current  passing  from  or  to  a  grid  through  the  vacuous 
space. 

Grid  Voltage.  Grid  voltage  is  the  voltage  between  a  grid  and  a  specified  point  of  the 
cathode. 

Grid  Bias.     Grid  bias  is  the  direct  component  of  grid  voltage. 

Grid  Driving  Power.  Grid  driving  power  is  the  integral  of  the  product  of  the  instan- 
taneous values  of  the  alternating  components  of  the  grid  current  and  voltage  over  a  com- 
plete cycle. 

Anode  Current.  (Plate  Current.)  Anode  current  is  the  current  passing  to  or  from  an 
anode  through  the  vacuous  space. 

Anode  Voltage.  (Plate  Voltage.)  Anode  voltage  is  the  voltage  between  an  anode  and 
a  specified  point  of  the  cathode. 

Peak  (or  Crest)  Forward  Anode  Voltage.  Peak  (or  crest)  forward  anode  voltage  is  the 
maximum  instantaneous  anode  voltage  in  the  direction  in  which  the  tube  is  designed  to 
pass  current. 

Peak  (or  Crest)  Inverse  Anode  Voltage.  The  peak  (or  crest)  inverse  anode  voltage  is 
the  maximum  instantaneous  anode  voltage  in  the  direction  opposite  to  that  in  which  the 
tube  is  designed  to  pass  current. 

Tube  Voltage  Drop.  Tube  voltage  drop  in  a  gas-  or  vapor-filled  tube  is  the  anode  volt- 
age during  the  conducting  period. 

Anode  Dissipation.  Anode  dissipation  is  the  power  dissipated  in  the  form  of  heat  by  an 
anode  as  a  result  of  electron  and/or  ion  bombardment. 

Gas  Current.  A  gas  current  is  a  current  flowing  to  an  electrode  and  composed  of  pos- 
itive ions  which  have  been  produced  as  a  result  of  gas  ionization  by  an  electron  current 
flowing  between  other  electrodes. 

Leakage  Current.  A  leakage  current  is  a  current  which  flows  between  two  or  more 
electrodes  by  any  other  path  than  across  the  vacuous  space. 

Electrode  Conductance.  (Variational.)  Electrode  conductance  is  the  ratio  of  the  in- 
phase  component  of  the  electrode  alternating  current  to  the  electrode  alternating  voltage, 
all  other  electrode  voltages  being  maintained  constant. 

NOTE:  As  most  precisely  used,  the  term  refers  to  infinitesimal  amplitudes. 

Electrode  Resistance.  (Variational.)  Electrode  resistance  is  the  reciprocal  of  the 
electrode  conductance. 

Electrode  Admittance.  Electrode  admittance  is  the  ratio  of  the  alternating  component 
of  the  electrode  current  to  the  alternating  component  of  the  electrode  voltage,  all  other 
electrode  voltages  being  maintained  constant. 

NOTE:  As  most  precisely  used,  the  term  refers  to  infinitesimal  amplitudes. 

Electrode  Impedance.    Electrode  impedance  is  the  reciprocal  of  the  electrode  admittance. 

Transadmittance.  Transadmittance  between  two  electrodes  is  the  ratio  of  the  alter- 
nating component  of  the  current  of  one  electrode  to  the  alternating  component  of  the 
voltage  of  the  other  electrode,  all  other  electrode  voltages  being  maintained  constant. 

NOTE:  As  most  precisely  used,  the  term  refers  to  infinitesimal  amplitudes. 

Transconductance.  Transconductance  between  two  electrodes  is  the  ratio  of  the  in- 
phase  component  of  the  alternating  current  of  one  electrode  to  the  alternating  voltage  of 
the  other  electrode,  all  other  electrode  voltages  being  maintained  constant. 

NOTE:  As  most  precisely  used,  the  term  refers  to  infinitesimal  amplitudes. 


4-06  ELECTRON  TUBES 

Control-grid — Plate    Transconductance.     (Transconductance,    Mutual    Conductance.) 

Control-grid — plate  transconductance  is  the  name  for  the  plate-current  to  control-grid 
voltage  transconductance. 

Conversion  Transconductance.  Conversion  transconductance  is  the  ratio  of  the  magni- 
tude of  a  single  beat-frequency  component  (/i  +  fz)  or  (/i  —  /2) ,  of  the  output  electrode 
current  to  the  magnitude  of  the  control  electrode  voltage  of  frequency  /i  under  the 
conditions  that  all  direct  electrode  voltages  and  the  magnitude  of  the  electrode  alternat- 
ing voltage  /2  remain  constant. 

NOTE:  As  most  precisely  used,  the  term  refers  to  an  infinitesimal  magnitude  of  the  voltage  of  fre- 
quency /i. 

Mu  Factor.  The  mu  factor  is  the  ratio  of  the  change  in  one  electrode  voltage  to  the 
change  in  another  electrode  voltage,  under  the  conditions  that  a  specified  current  remains 
unchanged  and  that  all  other  electrode  voltages  are  maintained  constant.  It  is  a  measure 
of  the  relative  effect  of  the  voltages  of  two  electrodes  on  the  current  in  the  circuit  of  any 
specified  electrode. 

NOTE:  As  most  precisely  used,  the  term  refers  to  infinitesimal  changes. 

Amplification  Factor.  The  amplification  factor  is  the  ratio  of  the  change  in  plate  volt- 
age to  a  change  in  control-electrode  voltage,  under  the  conditions  that  the  plate  current 
remains  unchanged  and  that  all  other  electrode  voltages  are  maintained  constant.  It  is  a 
measure  of  the  effectiveness  of  the  control-electrode  voltage  relative  to  that  of  the  plate 
voltage  on  the  plate  current.  The  sense  is  usually  taken  as  positive  when  the  voltages  are 
changed  in  opposite  directions. 

NOTE:  As  most  precisely  used,  the  term  refers  to  infinitesimal  changes.    See  also  Mu  Factor. 

Electrode  Characteristic*  An  electrode  characteristic  is  a  relation,  usually  shown  by  a 
graph,  between  an  electrode  voltage  and  current,  other  electrode  voltages  being  maintained 
constant. 

Transfer  Characteristic.  A  transfer  characteristic  is  a  relation,  usually  shown  by  a 
graph,  between  one  electrode  voltage  and  another  electrode  current. 

Inter  electrode  Capacitance.  Interelectrode  capacitance  is  the  direct  capacitance  be- 
tween two  electrodes. 

Electrode  Capacitance.  Electrode  capacitance  is  the  capacitance  of  one  electrode  to  all 
other  electrodes  connected  together. 

Input  Capacitance.  The  input  capacitance  of  a  vacuum  tube  is  the  sum  of  the  direct 
capacitances  between  the  control  grid  and  cathode  and  such  other  electrodes  as  are  oper- 
ated at  the  alternating  potential  of  the  cathode. 

Output  Capacitance.     The  output  capacitance  of  a  vacuum  tube  is  the  sum  of  the 

direct  capacitances  between  the  output 
electrode  (usually  the  plate)  and  the 
cathode  and  such  other  electrodes  as  are 
operated  at  the  alternating  potential  of 
the  cathode. 

QUIESCENT  POINT  AND  OPERAT- 
ING RANGE.  The  quiescent  point  is  the 
point  on  the  plate  characteristic  that 
represents  operating  conditions  with  no 
signal  applied  to  the  grid.  With  a  load  in 
the  plate  circuit,  it  may  be  determined  as 
follows.  For  a  load  resistance  of  r  ohms 
the  slope  of  the  load  characteristic  line  is 

E&o      i      ET65  2  1/r  =  —  AI/AJ!?,  drawn  from  the  point 

Plate  Voltage  Ebb  (Fig.   1)   corresponding  to  the  plate 

FIG.  1.    Tube  and  Load  Characteristics  supply  voltage.     The  intersection   y  of 

the  two  characteristic  curves  is  the  quies- 
cent operating  point,  giving  a  plate  voltage  of  EIO  and  a  plate  current  of  7&0. 

The  plate  supply  voltage  Ebb  is  divided  into  the  voltage  drop  across  the  tube,  Et,0, 
and  the  voltage  drop  across  the  load,  Ebb  —  Ebo- 

When  the  plate  supply  voltage  is  varied  over  the  range  1  to  2,  the  load  resistance  line 
will  shift  parallel  to  the  position  shown  and  over  the  operating  range  1  to  2.  The  inter- 
sections of  the  load  line  with  the  tube  characteristic  determine  the  corresponding  variation 
in  Ebo  and  Ibo* 

WTien  the  plate  supply  voltage  is  constant  but  the  tube  characteristic  is  changed, 
for  example  by  changing  the  grid  voltage,  the  intersection  of  the  characteristics  for  the 
changed  values  of  grid  voltage  with  the  load  line  determines  the  change  in  Ibo  and  Ebo- 


£ 


DEFINITIONS 


4-07 


In  Fig.  1  a  voltage  in  the  grid  circuit  changes  the  characteristic  curve  so  that  it  intersects 
the  load  line  at  points  3  or  4.  The  projections  of  these  points  on  the  two  axes  show  the 
change  in  I  bo  and  Ebo- 

The  tube  characteristic,  unlike  the  load  characteristic,  cannot  be  shifted  parallel  to 
the  initial  position  to  represent  other  operating  conditions,  since  it  usually  changes  shape. 
The  tube  characteristics  should  be  known  for  a  few  voltages  throughout  the  range  of 
operation.  Intermediate  values  may  be  interpolated. 

A-C  EQUIVALENT  CIRCUIT.  In  the  circuit  of  Fig.  2  the  a-c  voltage  Eg  in  the  grid 
circuit  produces  an  a-c  plate  current  Ip  and  an  a-c  plate  voltage  Ep.  For  small  a-c  voltages 


FIG.  2.     Triode  with  Resistance  Load 


FIG.  3.     Equivalent  A-c  Circuit 
of  Tube  with  Resistance  Load 


the  tube  is  equivalent  to  a  generator  with  an  internally  generated  voltage  pEe  and  in- 
ternal resistance  rp.    Figure  3  is  the  a-c  equivalent  of  the  circuit  in  Fig.  2. 
From  Fig.  3  the  a-c  plate  current  is 


The  a-c  plate  voltage  is 


The  voltage  amplification  is 


E, 


rP  -f-  r 
Ipr  - 


g  -— 

Tp  -J—  / 


The  power  output  of  the  tube  is 


'—••>- 


£& 


(4) 


Figure  4  illustrates  an  a-c  equivalent  of  the  circuit  of  Fig.  2  in  which  the  tube  is  repre- 
sented as  a  generator  of  constant  current  I  =  gm  Eg. 
The  a-c  plate  voltage  is 


The  a-c  plate  current  is 


(7) 
(8) 


The  constant-current  form  of  representation  is  convenient  for  calculation  when  the 
load  consists  of  a  number  of  parallel  elements  or  when  the  plate  resistance  of  the  tube  is 
high  and  rp/(rp  +  r)  approaches  unity. 


FIG.  4.    Equivalent  A-c  Circuit 
of  Tube  with  Resistance  Load. 


FIG.  5.    Equivalent  Circuit  of  Triode 


When  the  tube  is  amplifying  a-c  voltages  at  frequencies  at  which  the  capacitances  of 
tube  electrodes,  socket,  arid  wiring  are  not  negligible  the  equivalent  a-c  circuit  is  as  shown 
in  Fig.  5. 

The  capacitances  marked  Cgp,  Cgk,  Cpk  represent  the  grid-plate,  grid-cathode,  and 
plate-cathode  capacitances  (see  Section  5,  Article  25)  .  Any  socket  and  wiring  capacitances 


4-08 


ELECTRON  TUBES 


can  be  added  in  parallel.  In  screen-grid  and  other  multielectrode  tubes  where  the  addi- 
tional electrodes  are  grounded  the  circuit  reduces  to  an  equivalent-triode  network  similar  to 
Fig.  3. 

BALLAST  TUBES  AND  VOLTAGE  REGULATORS.  A  ballast  tube  is  used  as  a  series 
resistance  to  limit  the  load  current.  It  is  designed  for  a  definite  current  and  voltage  drop. 
Over  the  useful  portion  of  its  characteristic  a  large  change  in  voltage  accompanies  a 
small  change  hi  current.  As  a  result  of  this  characteristic  a  large  part  of  any  line-voltage 
change  is  absorbed  by  the  ballast  tube  and  a  relatively  small  change  occurs  at  the  load. 

In  reading  data  for  the  characteristic  curves,  it  may  be  necessary  to  allow  a  few  minutes 
after  each  change  in  voltage  for  the  tube  to  reach  its  temperature  equilibrium  condition. 

The  performance  may  be  determined  graphically  by  a  method  similar  to  that  described 
for  determining  the  quiescent  operating  point  by  plotting  the  line  for  a  resistance  load 
with  the  experimentally  determined  current-voltage  characteristic  of  the  ballast  tube. 

A  voltage-regulator  tube  is  operated  in  parallel  with  the  load.  Its  characteristic  shows 
a  large  change  in  current  for  a  small  change  in  voltage  near  the  operating  point.  When 
connected  in  parallel  with  the  load  with  a  suitable  resistance  effective  in  the  supply  voltage 
source  any  change  in  the  supply  voltage  will  cause  a  change  in  the  current  in  the  voltage- 
regulator  tube  such  that  the  voltage  across  the  regulator  tube  and  load  remains  practically 
constant. 

The  voltage-regulator  tube  is  designed  for  a  definite  voltage  and  is  operated  between 
specified  minimurn  and  maximum  current  limits.  A  certain  starting  voltage  somewhat 
higher  than  the  operating  voltage  is  required. 

The  operating  characteristics  are  most  conveniently  obtained  in  a  normal  operating 
circuit. 


4.  METHODS  OF  MEASURING  TUBE  CURRENTS  AND  PARAMETERS 

The  characteristic  relations  between  the  direct  voltages  and  currents  of  the  electrodes 
of  a  tube  may  be  obtained  in  a  static-characteristic  measuring  circuit  arranged  as  in  Fig.  6. 

The  voltages  applied  to  the 
different  electrodes  as  illus- 
trated are  measured  from  a 
unipotential  cathode.  If  the 
tube  has  &  filamentary  cathode 
it  is  understood  that,  when 
operating  with  direct-current 
filament  supply  such  as  when 
measuring  static  character- 
istics, the  electrode  voltages 
are  measured  from  the  neg- 
ative filament  terminal. 
With  alternating-current  op- 
eration of  a  filamentary 
cathode,  the  center  of  the 
filament  is  used  as  the  da- 


FIG.  6.     Circuit  for  Measuring  Static  Characteristics 


turn  of  potential  and  the  electrode  voltages  are  corrected  for  one-half  of  the  filament 
voltage.  Ordinarily  only  the  control-grid  bias  voltage  is  made  more  negative  by  approxi- 
mately one-half  of  the  peak  alternating  voltage  on  the  filament. 

FILAMENT  OR  HEATER  CHARACTERISTIC.  The  filament  or  heater  current  is 
obtained  for  several  values  of  filament  or  heater  voltage  ranging  from  values  producing 
temperatures  too  low  to  give  appreciable  electron  emission  to  values  producing  the  maxi- 
mum safe  operating  temperature.  The  other  electrodes  should  be  at  zero  voltage. 

The  curves  are  plotted  with  filament  or  heater  voltage  as  abscissas  and  filament  or 
heater  current  and  power  as  ordinates. 

CATHODE  HEATING  TIME.  The  cathode  heating  time  is  defined  for  purposes  of 
measurement  as  the  interval  from  the  time  of  application  of  filament  or  heater  voltage 
to  the  time  at  which  the  rate  of  increase  of  plate  current  is  a  maximum. 

If  the  primary  winding  of  a  transformer  is  connected  in  the  plate  circuit  and  a  meter 
is  connected  to  the  secondary  winding,  then  the  instant  at  which  the  rate  of  increase  of 
plate  current  is  a  maximum  is  indicated  by  a  maximum  reading  on  the  meter. 

The  voltage  at  the  terminals  of  the  filament  or  heater  should  remain  constant  at  the 
rated  or  specified  value. 

Because  of  the  very  slow  rate  of  change  of  plate  current  in  the  usual  tube,  the  charac- 
terisjtaes  of  the  transformer  or  meter  do  not  greatly  influence  the  result.  A  step-down 


METHODS  OF  MEASURING  TUBE  PARAMETERS 


4-09 


transformer  and  a  current  meter  which  is  sufficiently  damped  though  not  too  sluggish 
.are  suitable. 

EMISSION  CHARACTERISTIC.  The  emission  characteristic  shows  the  emission  cur- 
rent plotted  as  a  function  of  the  cathode  heating  power. 

The  readings  are  obtained  with  all  electrodes,  except  the  cathode,  connected  together 
as  the  anode  and  with  sumcient  positive  voltage  applied  to  the  electrodes  to  draw  the 
entire  emission  current  from  the  cathode.  Since  the  emission  current  at  normal  filament 
power  may  be  so  great  as  to  damage  the  tube,  readings  are  taken  at  lower  filament  powers 


0000.0 


ooo.o 


00.0 


-p> 


s=sss  °g 


FIG.  7.     Emission  Characteristic 


only,  and  normal  emission  current  is  obtained  by  extrapolation.  A  suitable  procedure  is 
as  follows,  the  values  applying  to  ordinary  receiving  tubes.  Readings  of  cathode  heating 
power  are  taken  with  emission  currents  of  0.1,  0.2,  0.5,  1.0,  2.0,  and  5.0  ma,  with  45  volts 
positive  applied  to"  the  anode.  The  results  are  plotted  in  Davisson  coordinates  (see  Fig.  7), 
which  are  a  special  system  of  curvilinear  coordinates.  If  the  emission  follows  Richardson's 
temperature  law  and  the  cathode  cooling  follows  the  Stefan-Boltzmann  law  of  radiation,  the 
characteristic  will  be  a  straight  line  when  plotted  in  these  coordinates.  The  observed 
points  may  be  extended  or  extrapolated  to  obtain  the  emission  at  normal  filament  power. 

The  emission  characteristic  for  a  type-80  tube  plotted  according  to  the  above  procedure 
is  shown  in  Fig.  7. 

If  the  curve  of  the  experimental  data  plotted  in  Davisson  coordinates  is  not  a  straight 
line  this  may  be  due  to  one  or  more  of  the  following  conditions: 

1.  Departure  from  the  Stefan-Boltzmann  cooling  (bends  downward). 

2.  Anode  voltage  too  low  to  draw  off  all  the  electrons  (bends  downward). 


4-10  ELECTRON  TUBES 

3.  Effect  of  cooling  due  to  heat  of  evaporation  of  electrons  (bends  downward).    The 
cooling  due  to  electron  evaporation  amounts  to  approximately  $IS  watts,  where  Is  repre- 
sents the  emission  current  in  amperes  and  <f>  represents  the  work  function  of  the  cathode 
in  volts.     This  effect  may  be  considerable  in  transmitting  tubes  where  the  currents  are 
high  and  in  tungsten-filament  tubes  where  the  work  function  is  large. 

4.  Poor  vacuum  (gas  ionization  effects)  (bends  upward) . 

5.  Heating  of  the  anode  by  the  emission  current  (bends  upward). 

6.  Progressive  change  in  activity  of  the  cathode. 

Reliable  analytical  data  cannot  be  obtained  by  this  method  when  these  extraneous 
effects  are  appreciable. 

ELECTRON  EMISSION.  Normal  electron  emission  is  determined  with  the  filament 
voltage  adjusted  to  the  normal  rated  value. 

All  electrodes  in  the  tube,  except  the  cathode  and  heater,  are  connected  together,  and 
a  sufficiently  positive  voltage  with  respect  to  the  cathode  is  applied  to  them  to  obtain 
practically  the  full  electron  emission. 

For  power-type  tubes  this  test  is  not  advisable  on  account  of  possible  damage  to  the 
tube.  The  method  of  extrapolation  described  under  Emission  Characteristic  should  be 
used. 

For  receiving-type  tubes  a  check  on  the  emissive  condition  of  the  cathode  can  usually 
be  made  safely  if  the  time  of  application  of  the  voltage  is  not  permitted  to  exceed  that 
required  for  rapid  reading  of  the  emission  current  meter.  An  anode  voltage  of  about  45 
volts  is  used. 

Since  this  test  usually  results  in  the  liberation  of  gas  and  abnormal  heating  of  the 
electrodes,  it  should  be  postponed  until  after  the  completion  of  other  tests,  or  a  sufficient 
time  should  elapse  between  this  and  other  tests  for  clean-up  and  return  to  normal  tempera- 
ture conditions. 

For  tubes  with  extremely  low  cathode  heating  poiuer,  such  as  the  oxide-coated  low-filament- 
current  types  1R5,  1S4,  1T4,  etc.,  this  test  is  neither  reliable  nor  safe  for  the  tube.  In 
checking  the  emission  of  tubes  of  this  type  a  low  filament  voltage  is  applied  and  gradually 
increased  until  a  specified  emission  (less  than  normal  for  the  particular  type  of  tube, 
usually  3  to  5  ma)  is  obtained.  The  filament  voltage  required  to  obtain  the  specified 
emission  is  an  indirect  measure  of  the  cathode  or  filament  activity.  This  is  an  arbitrary 
method  suitable  for  comparing  tubes  of  the  same  type. 

In  general  a  safe  method  for  reading  emission  under  all  different  conditions  consists  in 
using  a  rotating  contactor  to  apply  the  voltage  for  only  a  small  fraction  of  the  time.  An 
oscillograph  is  used  to  read  the  emission  current  which  flows  during  the  small  interval 
of  time  that  the  voltage  is  applied. 

GRID  CHARACTERISTIC.  A  grid  characteristic  curve  shows  the  current  in  a  grid 
electrode  as  a  function  of  the  voltage  on  this  electrode.  The  voltage  on  all  the  remaining 
electrodes  is  held  constant.  A  family  of  curves  is  obtained  by  using  a  different  value  of 
voltage  on  one  of  the  remaining  electrodes  for  each  curve. 

The  grid  characteristic  used  most  frequently  is  that  of  the  control  grid  with  the  plate 
voltage  as  the  parameter  for  the  several  curves. 

In  reading  data  for  the  curves  the  current  should  not  be  allowed  to  flow  long  enough  to 
cause  abnormal  heating  of  the  grid.  The  readings  should  be  taken  near  enough  together  to 
show  any  irregularities  due  to  secondary  emission  or  gas. 

IONIZATION,  LEAKAGE,  AND  STRAY  EMISSION  CURRENTS.  In  vacuum  tubes 
the  normally  small  currents  due  to  ionization,  leakage,  and  electron  emission  from  electrodes 
other  than  the  cathode,  although  usually  negligible  in  the  plate  and  other  current-carrying 
electrodes,  may  have  an  appreciable  effect  in  the  control-grid  circuit  of  the  tube. 

The  total  current  flowing  to  the  negatively  biased  control  grid  may  be  divided  into  com- 
ponents as  follows: 

1.  Electrons  from  the  cathode  which  reach  the  grid  by  virtue  of  contact  potential  and 
initial  velocities. 

2.  Electrons  from  other  electrodes  to  the  control  grid. 

3.  Ionization  current. 

4.  Leakage  current. 

5.  Electron  emission  from  the  control  grid. 

Figure  8  illustrates  the  contributions  of  the  various  sources  enumerated  with  the  excep- 
tion of  2,  which  is  generally  negligible. 

The  several  components  may  be  separated  and  measured  by  the  following  methods. 

The  leakage  current  is  measured  with  a  direct  voltage  applied  between  any  two  elec- 
trodes and  without  any  connections  on  the  other  electrodes.  The  tube  should  be  operated 
with  normal  voltages  and  currents  until  all  parts  have  reached  full  operating  temperature. 
The  filament  voltage  is  then  disconnected  and  the  leakage  currents  read  while  the  insulation 


METHODS    OF  MEASURING  TUBE  PARAMETERS         4-11 


•He 


is  at  normal  operating  temperature.  The  tube  should  be  complete  with  its  base  but 
without  socket  or  holder.  The  test  voltage  should  be  specified.  Normal  maximum  op- 
erating voltage  is  preferable. 

If  any  of  the  electrodes  remain  hot  enough  to  emit  electrons  an  error  will  be  introduced 
into  the  leakage  readings. 

The  grid  emission  can  be  measured  by  noting  the  current  at  a  bias  sufficiently  negative 
(point  A  in  the  figure)  to  stop  the  plate  current,  since  at  this  point  (A)  the  ionization 
current  (3) ,  being  proportional  to  the  plate  cur- 
rent, is  negligible. 

The  grid  emission  is  found  by  subtracting  the 
leakage  current  from  the  grid  current  at  the  point 
A.  If  the  leakage  current  is  negligible  the  test 
gives  the  grid  emission  directly. 

A  direct  measurement  of  grid  emission  can  be 
made  (when  leakage  current  is  negligible)  by 
connecting  the  test  voltage  between  the  grid  and 
plate  without  any  connection  to  the  cathode. 
The  positive  voltage  on  the  plate  draws  the  elec- 
trons emitted  by  the  grid.  The  cathode  should 
be  at  its  normal  operating  temperature.  The 
tube  should  be  operated  with  normal  operating 
voltages  for  a  time  preceding  this  test,  and  then 
quickly  switched  to  the  emission  test  circuit  and 
grid  emission  noted  while  the  electrodes  are  still 
approximately  at  their  normal  temperatures. 

The  ionization  current  (3)  is  the  difference  be- 
tween the  total  grid  current  and  the  sum  of  the 
leakage  (4)  and  emission  (5)  currents  in  the  range 
of  grid  bias  over  which  this  difference  is  propor- 
tional to  the  plate  current.  Departure  from  this 
proportionality  indicates  the  start  of  electron 
current  (1)  to  the  grid. 

PLATE  CHARACTERISTIC.  The  plate  char- 
acteristic gives  the  plate  current  as  a  function  of 
the  plate  voltage,  the  voltages  on  the  other  elec- 
trodes being  held  constant.  A  family  of  curves  may  be  obtained  by  using  a  series  of  volt- 
ages on  one  of  the  other  electrodes.  A  series  of  control-grid  voltages  is  ordinarily  used 
for  the  different  curves.  For  examples  see  Figs.  21,  24,  and  27. 

The  data  for  the  curves  are  read  in  the  static  characteristic  test  circuit.  For  the  range 
of  currents  and  voltages  beyond  the  normal  average  values,  it  is  sometimes  necessary  to 
employ  a  method  which  is  rapid  enough  to  avoid  heating  of  the  electrodes.  When  a  well- 
regulated  voltage  source  is  available  the  voltages  may  be  set  to  the  desired  values  and  a 
switch  closed  only  long  enough  for  rapidly  reading  the  current  on  a  suitably  damped  meter. 

GRID-PLATE  CHARACTERISTIC.  The  grid-plate  characteristic  or  transfer  character- 
istic gives  the  plate  current  vs.  grid  voltage  for  the  condition  of  constant  voltage  on  the 
remaining  electrodes.  Such  a  transfer  characteristic  may  be  taken  for  any  of  the  grids 
of  a  multigrid  tube.  The  curves  generally  used  show  the  control-grid  voltage  as  abscissa 
and  plate  current  as  ordinate,  several  curves  being  plotted  each  for  a  different  plate  voltage. 
See  for  examples  Figs,  22,  25,  and  29. 

The  data  for  the  curves  are  obtained  in  the  static-characteristic  test  circuit,  the  same 
precautions  being  observed  as  in  reading  data  for  plate  characteristics. 

CONDUCTANCE.  The  conductance  of  an  electrode  may  be  obtained  from  the  char- 
acteristic curve  showing  the  electrode  current  vs.  the  electrode  voltage.  The  slope  of 
this  curve  at  any  point  gives  the  electrode  conductance  at  the  voltages  represented  by 
the  point.  The  accuracy  of  the  measurement  as  determined  in  this  way  from  the  static 
characteristics  may  be  made  as  good  as  desired  by  reading  small  current  and  voltage  incre- 
ments with  sufficient  accuracy. 

The  electrode  resistance  is  the  reciprocal  of  the  conductance.  For  example,  the  plate 
resistance  is  the  reciprocal  of  the  plate  conductance,  the  grid  resistance  is  the  reciprocal 
of  the  grid  conductance,  etc. 

When  many  readings  are  to  be  made,  a  method  of  direct  'measurement  is  most  convenient. 
One  means  might  be  to  re,ad  the  alternating  current  produced  by  a  small  alternating 
voltage  in  the  circuit.  In  general  a  Wheatstone  bridge  circuit  is  preferred  as  shown  in  Fig.  9. 

In  this  circuit  a  small  alternating  voltage  (about  0.5  volt,  1000  cps)  is  applied  to  ter- 
minals 1-3  of  the  bridge.  The  electrodes  being  measured  are  connected  to  terminals  1-2, 


FIG.  8.     Stray  Electron  Currents 


4-12 


ELECTRON  TUBES 


and  the  bridge  is  balanced  with  r%  for  minimum  sound  in  the  telephone  receivers.    The  con- 
ductance of  the  electrode  j  is  given  by 

"-Si 

If  the  receivers  are  preceded  by  an  amplifier  the  alternating  input  voltage  may  be  kept 
small  enough  so  that  the  measurements  are  independent  of  the  magnitude  of  this  voltage. 


.  Voltages 
/Adjusted 

to  Test 

Conditions 


FIG.  9.     Conductance  Measurement  Circuit 


This  also  reduces  any  narmonics  produced  by  non-linearity  of  the  characteristic.  The 
stray  capacitances  of  the  wiring  and  parts  of  the  bridge  should  be  kept  low  or  balanced 
by  a  suitable  arrangement  of  parts.  A  variable  capacitance  C\  connected  across  r\  can  be 
used  to  balance  different  tube  capacitances. 

The  voltage  drop  in  the  bridge  circuit  should  be  taken  into  account  in  determining  the 
electrode  voltage  of  the  tube.  The  inductance  L  provides  a  low  d-c  resistance  path  across 
the  amplifier.  The  blocking  condensers  C  keep  the  direct  current  in  a  path  of  constant 
resistance  which  simplifies  the  correction. 

TRANSCONDTJCTANCE.  The  transconductance  is  a  measure  of  the  change  in  current 
in  one  electrode  produced  by  a  voltage  on  another  electrode.  In  a  tube  having  several 
electrodes  the  transconductance  may  be  measured  between  any  two  electrode  circuits, 


FIG.  10.     Transconductance  Measurement  Circuit 


the  cathode  being  common  to  both  circuits.  For  example,  the  grid-plate  transconductance, 
or  mutual  conductance,  may  be  determined  from  the  curves  for  the  grid-plate  transfer  char- 
acteristic since  this  shows  the  plate  current  vs.  grid  voltage.  The  slope  of  this  curve  at  a 
certain  point  gives  the  transconductance  at  the  voltages  represented  by  the  point. 

The  transconductance  may  be  measured  directly  by  means  of  the  circuit  of  Fig.  10.  An 
alternating  voltage  as  small  as  convenient  is  applied  in  this  circuit,  and  the  resistor  r\  is 
adjusted  for  minimum  sound  in  the  receivers.  The  stray  capacitances  due  to  the  wiring 


METHODS  OF  MEASURING  TUBE  PARAMETERS 


4-13 


and  the  tube  may  be  balanced  by  means  of  the  condenser  C  connected  either  between  points 
X-Y  or  X-Z  as  determined  by  trial.  The  direct  voltages  are  supplied  to  the  electrodes 
through  the  choke  coils  Li  and  L%.  The  reactance  of  the  choke  coils  at  the  frequency 
of  the  applied  alternating  voltage  should  be  large  with  respect  to  r%  and  r3.  If  the  resis- 
tances of  the  choke  coils  cause  an  appreciable  drop  in  direct  voltage  the  electrode  voltages 
should  be  corrected  accordingly. 

At  balance  the  transconductance  is  given  by 

«» =  £,  (10> 

This  relation  assumes  that  the  electrode  conductance  is  negligible.  If  the  conductance 
of  electrode  j  is  l/r?-  and  that  of  k  is  1/r^  the  transconductance  is  given  by 

ft._£fr*±«>A±j£  (11) 

The  following  circuit  constants  will  cover  a  range  of  transconductance  from  1  micromho 
to  10,000  micromhos. 

n  =  10  X      0.1  ohm          r2  =  1000  ohms 
10  X       1.    ohm 

10  X     10.    ohms         n  =  100  ohms 
10  X  100.    ohms 

With  Li  equal  to  50  henrys  and  La  equal  to  10  henrys  and  a  1000  cps  alternating  voltage 
the  error  in  using  the  simplified  equation  will  be  less  than  2  per  cent  if  ry  and  rjb  are  greater 
than  10,000  ohms  and  100,000  ohms,  respectively. 

MU  FACTOR.  The  mu  factor  is  the  ratio  of  the  voltages  in  two  electrode  circuits  re- 
quired to  maintain  constant  current  in  the  circuit  of  any  specified  electrode.  It  may 
be  determined  from  the  static  characteristic  curves  or  measured  by  a  balance  method. 
For  example,  the  amplification  factor  is  a  special  case  in  which  the  control-grid  voltage 
and  plate  voltage  are  changed  in  such  a  way  as  to  maintain  constant  plate  current. 

In  the  circuit  of  Fig.  1 1  the  electrode  in  which  the  current  is  to  be  held  constant  is 
connected  to  point  A.  The  other  two  electrodes  entering  directly  in  the  measurement 


D.CX 

Voltages, 
>  Adjusted 
'  to  Testr 

Conditions 


FIG.  11.     Mu  Factor  Measurement  Circuit 

are  connected  to  points  B  and  C,    When  r\  and  ra  and  M  are  adjusted  for  minimum  sound 
in  the  telephone  receivers,  the  mu  factor  is  given  by 


!•? 

n 


The  direct  voltages  to  the  electrodes  under  test  may  be  supplied  through  choke  coils  L 
having  a  reactance  at  the  frequency  of  the  alternating  voltage  which  is  high  with  respect  to 


4-14 


ELECTRON  TUBES 


the  resistances  n,  ra,  and  rs.  The  resistances  of  the  chokes  should  be  low  enough  to  cause 
negligible  loss  in  direct  voltage  or  the  true  electrode  voltage  determined  by  subtracting  the 
loss.  If  the  electrode  conductances  are  not  negligible  in  comparison  with  n,  r2>  and  rs  a 
correction  should  be  made  for  this. 

INTERELECTRODE  CAPACITANCE.  The  capacitance  between  the  electrodes  of  a  tube 
may  be  measured  in  various  ways.  It  is  preferable  to  read  the  direct  capacitance  between 
any  two  electrodes  rather  than  the  total  capacitance  between  an  electrode  and  all  other 
electrodes.  The  readings  are  normally  made  with  the  tube  cold  and  no  direct  voltages 
applied.  When  the  tube  is  heated  the  capacitance  changes  a  small  amount  owing  to 
the  presence  of  space  charge,  but  the  change  is  ordinarily  negligible.  The  tube  should  be 

complete  with  base,  though  the 

A  socket  capacitance  is  not  included 

in  the  measurement.  For  most 
reliable  results,  readings  should 
not  be  taken  with  any  electrodes 
disconnected,  and  the  tube  should 
be  mounted  in  a  specified  way 
with  respect  to  any  shields.  For 
indirectly  heated  types  the  fila- 
ment and  cathode  should  be  tied 
together. 

A  bridge  method  for  the  meas- 
urement of  direct  interelectrode 
capacitance  in  a  triode  is  shown 
in  Fig.  12.  The  capacitance  to 
be  measured  is  connected  to  the 
terminals  A-B.  The  figure  shows 
the  grid-plate  capacitance  Cgp 
connected  for  measurement.  The 
effect  of  the  grid-cathode  capaci- 
tance Cgjo  in  the  circuit  across  r^ 
is  ordinarily  negligible  owing  to 
the  low  resistance  of  r^  The 
plate-cathode  capacitance  Cpk  is 
across  the  amplifier  and  tele- 
The  standard  capacitance  C  and 
When  the  bridge  is  bal- 


FIG.  12.    Electrode  Capacitance  Measurement  Circuit 


phone  receivers,  which  does  not  affect  the  balance, 
resistance  r  are  adjusted  for  minimum  sound  in  the  receivers, 
anced  the  capacitance  is 


(12) 


An  error  in  the  reading  may  result  if  appreciable  leakage  resistance  exists  across  Cx. 

5.  VACUUM-TUBE  OPERATION 

ELECTRON  TRANSIT  TIMES  AND  INERTIA  EFFECT.  Since  the  single  electron 
has  a  mass  of  9.035  X  10 ~28  gram  there  will  be  a  force  acting  on  the  electron  in  the  presence 
of  an  electric  field  which  is  proportional  to  the  product  of  the  field  strength  and  the  electron 
unit  o_f  charge.  This  force  will  accelerate  the  electron,  giving  it  finite  velocities.  If  the 
electron  has  started  from  rest  and  attains  final  velocities  much  smaller  than  the  speed  of 
light,  the  velocity  in  practical  units  may  be  expressed  as 

v  =  5.95  X  10  V?    cm  per  sec 

where  V  is  in  volts.    It  is  common  practice  to  express  the  velocity  of  an  electron  in  terms 
of  the  voltage  in  the  above  expression  instead  of  the  usual  centimeters  per  second. 

If,  however,  the  velocity  of  the  electron  has  been  accelerated  to  velocities  not  negligible 
compared  to  the  speed  of  light,  c,  which  is  3  X  1010  cm  per  sec,  the  mass  of  the  electron, 
77i,  will  increase  in  the  ratio 


—  =  1.96  X  10-T  +  [(1.96  X  10-6F)2  + 


(13) 


where  me  is  the  mass  of  the  electron  at  rest.    At  100,000  volts  the  increase  in  mass  is  about 
22  per  cent,  whereas  at  1000  volts  the  increase  is  only  about  0.2  per  cent.    At  very  high 


VACUUM-TUBE  OPERATION 


4-15 


voltages,  therefore,  the  velocity  of  an  electron  will  be  less  than  the  above  expression  by  the 
square  root  of  mjm,  the  resulting  velocity  being 


(14) 


=  5.95  X  107  Vy  —    cm  per  sec 

¥      m 


The  transit  time  of  an  electron  becomes  of  importance  at  very  high  frequencies  and  as 
would  be  expected  depends  upon  the  electron  velocity  expressed  at  some  convenient  point 
and  the  distance  it  has  traveled.  The  constant  K  in  the  following  expressions  is  a  factor 
taking  into  account  the  geometry  of  the  electric  field  and  the  effect  of  space  charge,  if 
present,  on  the  electric  field.  For  all  the  cases  where  the  electron  has  been  assumed  to 
start  with  zero  velocity  the  transit  time,  r,  is 


KD 


5.95  X  107 


X 


(15) 


where  D,  in  centimeters,  is  the  distance  traveled  from  rest  to  the  point  of  voltage  V,  in 
volts.  The  dimensionless  constant  K  has  the  following  values:  K  =  1  for  a  field-free  space; 
K  =  2  for  field  between  parallel  planes  without  space  charge;  K  =  3  for  field  between 
parallel  planes  with  space  charge. 

For  concentric  cylinders  the  values  of  K  are  given  in  Fig.  13.    The  A  and  B  curves  are 
for  the  usual  case  where  the  higher  voltage  is  on  the  outside  cylinder  (the  cathode  being 


7.0 
6.0 

5.0 

r 

*z 

3.0 
2.0 
1.0 

Transit  T 

me  Factor 
Space  Cha 
No  Space 

for  Cylin 
rge  Li  mi 
Charge 

d 
ec 

lea 
i  C 

1  Diodes 

^ 

-^ 



J 

s'* 

X 

•^ 

TQSSS  cathode  radius 
"*•=:  anode  radius      D=(r- 
Transit  time  = 
KD                 KD             1 

—it 

5 

sp 
no 

A^ 

^^ 

^X^ 

i-p  ^ 

x 

^ 

-x 

S> 

x^ 

^ 



•0)  cm. 

^^ 

-.-' 

,-' 

•^* 

c. 

^^^ 

„-• 

.- 

•» 

-* 

V/2^) 
\  m  ' 

Corollary:. 
For  parallel 
r/r0=l= 



5.95-1 
planes 

v- 

o1'  VE, 
{K=! 

^^ 

-"•" 

<^ 

*  — 

•—  *. 

••», 

•«. 

A 

ace  chg. 

*"*—  «.«^  _ 

-—  .- 

.__ 

—  •    — 



.— 

-,- 







1.  2.        3.     4.  5.  6.    8,  10  20       30    40       60    80  100          200     300       500    700    1000 

r0/r  for  Curves  A  &  B  ;  r/r0f°r  Curves  A  &  8 

FIG.  13.     Electron-transit  Time  for  an  Electron  Starting  with  Zero  Initial  Velocity.    Chart  Includes 
Cylindrical  and  Parallel  Plane  Structures  in  an  Electric  Field  with  and  without  Space  Charge.    (Cour- 
tesy RCA  Review.) 

the  inner  one) ;  the  Af  and  Bf  curves  are  for  the  case  where  the  positive  voltage  is  on  the 
inner  cylinder. 

In  many  cases  one  is  interested  in  the  transit  time  when  the  electron  is  not  starting  from 
rest,  as,  for  example,  for  the  electron  transit  from  the  control  grid  to  the  screen  grid  region 
of  a  pentode.  For  this  case 

sec  (16) 


5.95  X  107  ' 

where  K  and  Z>  are  defined  as  before,  Vi  and  Vz  are  the  voltages  of  the  two  points  between 
which  the  electron  is  passing,  and  Vz  >  Fi. 

THE  INPUT  ADMITTANCE  OF  VACUUM  TUBES.     The  input  admittance  of  nega- 
tive-grid-controlled vacuum  tubes  may  have  deleterious  effects  especially  at  high  fre- 


4-16  ELECTRON   TUBES 

quencies.  Its  conductive  component  is,  to  a  first  order,  proportional  to  the  square  of  the 
input-signal  frequency.  Being  a  power-absorbing  element  it  will  thus  broaden  the  fre- 
quency-response characteristic  of  the  input  circuit  and  lower  the  anticipated  gain  of  the 
preceding  stage.  Furthermore,  it  also  varies  with  the  tube  transconductance  or  tube  gain. 
The  reactive  component  of  admittance  is  fortunately  independent  of  frequency,  but  un- 
fortunately it  varies  with  the  tube  transconductance.  For  a  triode  or  a  pentode  radio- 
frequency  amplifier  the  effect  is  to  increase  the  input  capacitance  of  the  tube  in  the  order  of 
1  to  2  jtt/xf  as  the  tube  gain  is  varied  from  cutoff  to  its  maximum  value.  At  low  frequencies 
this  small  capacitance  change  is  made  negligible  by  padding  the  input  circuit  with  a  suf- 
ficiently large  fixed  capacitance,  but  at  high  frequencies  this  precaution  is  impractical. 
These  admittance  variations  which  change  with  the  tube  transconductance  may  be  greatly 
reduced  by  the  use  of  an  unbypassed  cathode  resistor  so  positioned  that  it  will  be  common 
to  both  the  input  and  output  circuits  of  the  tube. 

The  input  conductance,  gg,  of  a  vacuum  tube  consists  essentially  of  two  components. 
One  of  these  components,  gt,  has  its  origin  in  the  electron-transit  time  phenomena,  and  the 
other,  gLi  has  its  origin  in  that  portion  of  the  cathode-lead  inductance  which  is  common  to 
both  the  input  and  the  output  terminals  of  the  vacuum  tube.  Quantitatively  the  transit- 
time  loading,  gt,  as  determined  by  D.  O.  North  is 

ra  17  +  35  (rs/n)  20  fa/n)2 


where  gt  is  the  transit  time  input  loading  in  mhos,  gik  is  the  signal-grid-to-cathode  trans- 
conductance  in  mhos,  /  is  the  signal  frequency  in  cycles  per  second,  n  is  the  transit  time  in 
seconds  for  an  electron  to  move  from  the  cathode  to  the  plane  of  the  signal  grid,  T2  is  the 
transit  time  in  seconds  for  an  electron  to  move  from  the  signal  grid  plane  to  the  plate, 
and  -Dp  and  vg  are  the  electron  d-c  velocities  at  the  plate  and  grid  respectively.  It  should  be 
noted  that  for  a  well-screened  tetrode  or  pentode  the  transit  time,  ra,  and  the  plate  velocity, 
VP,  are  with  reference  to  the  screen  grid.  Also  it  should  be  noted  that  the  grid-to-cathode 
transconductance  of  a  triode  is  the  same  as  its  grid-to-plate  transconductance,  whereas  for 
a  pentode  or  tetrode  the  grid-to-cathode  transconductance  has  a  value  equal  to  the  grid-to- 
plate  transconductance  multiplied  by  the  ratio  of  cathode  current  to  plate  current.  The 
last  two  terms  in  the  brackets  of  the  above  expression  are  only  of  second  order  of  magnitude 
in  the  conventional  amplifier  tubes  since  the  ratio  vp/vg  is  approximately  equal  to  10.  In- 
spection of  this  expression  further  indicates  that  considerable  loading  may  be  contributed 
to  the  input  circuit  by  electrons  moving  in  the  space  between  the  control  grid  and  plate 
(or  screen  grid)  .  As  a  precaution  in  using  the  above  relationship,  the  simplifying  assump- 
tions applied  for  its  derivation  are  listed  below. 

1.  The  transit  times,  n  and  ra,  are  small  compared  to  the  period,  I//. 

2.  The  electrodes  are  parallel  planes. 

3.  The  initial  velocity  of  the  emitted  electrons  is  zero  and  the  emission  is  ample,  so  that 
the  three-halves  power  of  voltage  vs.  current  holds  in  the  cathode-grid  region. 

4.  The  amplification  factor  of  the  signal  grid  is  high,  so  that  electrons  on  one  side  do  not 
appreciably  influence  the  field  on  the  other  side. 

5.  The  grid  is  an  equipotential  plane  surface. 

6.  The  alternating  voltage  at  the  grid  is  very  small  with  respect  to  the  effective  static 
grid  potential. 

7.  The  alternating  voltage  at  the  plate  (or  screen  grid)  is  zero. 

8.  The  potential  between  grid  and  plate  (or  screen  grid)  is  substantially  linear  and  free 
of  space  charge. 

That  part  of  the  input  conductance,  gz,  which  stems  from  the  cathode-lead  inductance 
may  be  expressed  to  a  first  approximation,  as  shown  by  J.  0.  Strutt  and  Van  der  Ziel,  by 

gL  =  ^FLkCgkgik  (IS) 

where  gL  is  the  input  loading  due  to  cathode-lead  inductance  in  mhos,  /  is  the  signal  fre- 
quency in  cycles  per  second,  Lk  is  the  cathode-lead  inductance  common  to  both  input  and 
output  circuits  in  henrys,  Cgk  is  the  capacitance  between  the  signal  grid  and  the  cathode 
in  farads,  and  gik  is  the  signal-grid-to-cathode  transconductance  in  mhos. 

Since  both  the  aforementioned  components  of  input  loading  are  proportional  to  the  signal 
frequency  squared  and  to  the  tube  transconductance,  the  two  effects  are  therefore  im- 
practical to  measure  separately.  It  is  also  difficult  to  measure  the  cathode-lead  inductance. 
If  one  estimates  this  inductance  to  be  5  to  10  millimicrohenrys  for  the  glass  miniature  tubes 
and  10  to  15  millimicrohenrys  for  the  single-ended  metal  or  loctal  tubes  where  the  lower 
values  are  used  for  those  tubes  having  two  cathode  leads,  we  find  that  from  10  per  cent  to 


VACUUM-TUBE  OPEEATION 


4-17 


about  35  per  cent  of  the  total  input  loading  is  contributed  by  the  effect  of  cathode-lead 
inductance. 

The  screen-grid  lead  inductance,  L&,  in  a  pentode  amplifier  or  tetrode  amplifier  in- 
troduces negative  input  loading,  gi&,  which  quantitatively  is 

giA 47r2/2L4aC«i«2gi2  (19) 

where  /  is  the  frequency  in  cycles  per  second,  L&  is  the  screen-grid  lead  inductance  in 
henrys,  Cgigz  is  the  capacitance  between  the  signal  grid  and  the  screen  grid  in  farads,  and 


600 
500 

V) 

j:  400 
E 

=J. 
^-  300 

a 

-  200 
Q> 

100 

n 

/ 

5SJ7 

SSK: 

/ 

/ 

/ 

/ 

/ 

/  / 

// 

/ 

100 

|    so 

=t 

^.     60 

3 
0. 

»  4° 

20 
0 

3001 

jS 

c 

003 

X 

/ 

/ 

/ 

0  1000  2000  3000  "0  1000          2000          3000 

Transconductance,  jumhos  Transconductance,  /jmhos 


800 
S 

•|    600 
3. 
•5   400 

D. 

C 

c»  200 

r\ 

/ 

X^AB7 

~r>i 

^ 

^ 

"6SGT 

7 

.X 

V 

'S 

0  2000          4000         6000 

Transconductance,  //mhos 


g  Input,  jumhos 
10  £*  m  a 

0000 
3  0  0  0  O 

fifiJ    Z. 

5AU6 

^ 

'/ 

/, 

^ 

S 

^ 

0 

Transconductance, 


3500 
3000 

o2500 

£ 

=^2000 

J-1500 
&> 
1000 

500 


350 
300 
J250 
3,200 

g-150 
Co 
100 

50 
n 

e 
/ 

AG5 

/ 

/ 

/ 

/ 

/ 

f 

/ 

^ 

-   € 

AK5 

/ 

/ 

/^ 

S 

0  4000         8000        12000 

Transconductance,  fcmhos  Transconductance,  jumhos 

PIG.  14.     Input-loading  Conductance  vs.  Grid-plate  Transconductance  for  a  Group  of  R-f  Pentode 

Amplifying  Tubes  at  100  Me. 

£12  is  the  transconductance  from  signal  to  screen  grid.  This  negative  loading  is  small  in 
magnitude  compared  to  the  positive  loading  introduced  by  the  cathode-lead  inductance, 
and  for  the  typical  pentode  amplifier  (gu  »  0.2gi&,  Cgig2  <  Cgk,  and  L&  «  Lk)  it  will  reduce 
the  total  loading  by  only  a  few  per  cent.  If,  however,  additional  inductance  is  placed  in 
series  with  the  screen  lead  it  should  be  possible  to  neutralize  the  positive  input  loading 
completely,  with  the  distinct  disadvantage  that  instability  or  even  parasitic  osculations 
may  be  produced  in  the  amplifier  stage. 

If  the  static  tube  voltages  are  held  constant  so  that  the  electron-transit  times  do  not  vary 
and  the  frequency  only  is  varied,  we  may  write  the  total  input  loading,  gg,  as 

gg  =  gt  +  gL  =  gj*  l  (20) 

where,  if  the  g  terms  are  expressed  in  micromhos  and  the  frequency  /  in  megacycles,  the 
constant  K  will  be  in  the  unit  micromhos  per  megacycle  squared.  This  proportionality 
factor  K  is  listed  in  Table  1  for  several  tube  types  with  the  static  operating  voltages,  cur- 
rents, and  signal-grid-to-plate  transconductances  to  which  it  applies.  At  frequencies 
below  25  Me  a  conduction  term  negligible  at  higher  frequencies  should  be  added.  This  is 


4-18 


ELECTRON  TUBES 


due  to  dielectric  losses  and  is  proportional  to  frequency.    For  single-ended  metal  tubes  this 
added  conduction  loss,  gn,  is  approximately  equal  to 

gn  =  0.03/ 

when  /  is  in  megacycles  and  gn  is  in  micromhos.    For  the  glass  miniature  and  acorn  tubes 
this  loss  is  too  small  to  be  measured  accurately. 

It  should  be  noted  that  in  Table  1  the  five  entries  at  the  bottom  of  the  list  are  for  mixer  or 
converter  use.  Those  applications  where  the  signal  is  injected  on  an  outer  grid  and  the 
local  oscillator  is  injected  on  the  inner  grid  produce  negative  loading.  In  the  case  of  the 
6L7,  the  signal  being  placed  on  the  grid  adjacent  to  the  cathode,  positive  input  loading  is 
produced  as  in  the  amplifier  applications  discussed. 

If  the  grid-plate  transconductance  is  varied  from  minimum  bias  to  cutoff  bias  the  input 
loading  will  vary  in  proportion  to  this  transconductance  and  a  complicated  function  of  the 
electron  transit  times.  In  Fig.  14  are  given  in  graphical  form  some  measured  values  of  input 
loading.  These  data  were  observed  at  100  Me  with  the  static  voltages  of  Table  1  applied. 
The  grid-plate  transconductances  were  varied  by  changing  the  signal-grid  bias.  To  com- 
pute the  input  loading  at  any  frequency,  /  in  megacycles,  multiply  the  loading  at  100  Me 
by  the  ratio  (//100)2. 

Table  1.    Values  of  K,  Cgiet  and  AC  Input  for  Several  Tube  Types 


Grid-plate 

AC  Input 

Tube 
Type 

Plate 

Volt- 
age, 
volts 

Screen 
Volt- 
age, 
volts 

Signal- 
grid 
Bias, 
volts 

Plate 
Cur- 
rent, 
ma 

Screen 
Cur- 
rent, 
ma 

Transconduc- 
tance or  Con- 
version Trans- 
conductance, 

cgk* 

fj.p.t 

Change 
in  Input 
Capaci- 
tance^ 

Input 
Loading 
Constant 
K,  ^mhos 

pirnhos 

/i/if 

per  Me2 

6SJ7 

250 

100 

-3 

3.0 

0.8 

1650 

2.5 

1.0 

0.053 

6SK7 

250 

100 

-3 

9.2 

2.6 

2000 

2.1 

1.2 

0.050 

6SH7 

250 

150 

-  1 

10.8 

4.1 

4900 

4.2 

2.3 

0.063 

6SG7 

250 

125 

-1 

11.8 

4.4 

4700 

3.7 

2.3 

0.060 

6AB7 

300 

200 

-3 

12.5 

3.2 

5000 

3.6 

1.8 

0.079 

6AC7 

300 

150 

-2 

10.0 

2.5 

9000 

6.4 

2.4 

0.175 

9001 

250 

100 

-3 

2.0 

0.7 

1400 

1.7 

0.5 

0.0062 

9003 

250 

100 

-3 

6.7 

2.7 

1800 

1.4 

0.5 

0.0066 

6AK5 

150 

120 

-2 

7.5 

2.5 

5000 

2.6 

I.I 

0.0134 

6AG5 

250 

150 

-1,8 

7.0 

2.0 

5000 

3.9 

1.4 

0.033 

6BA6 

250 

100 

-1.2 

11.0 

4.2 

4400 

3.5 

2.2 

0.060 

6AU6 

250 

150 

-1 

10.8 

4.3 

5200 

3.5 

2.5 

0.076 

954 

250 

100 

-3 

2.0 

0.7 

1400 

1.5 

0.5 

0.005 

6J7 

250 

100 

-3 

2.0 

0.5 

1225 

2.5 

1.0 

0.05 

6K7 

250 

100 

-3 

7.0 

1.7 

1450 

2.1 

1.2 

0.05 

6A8 

250 

100 

-3 

3.5 

2.7 

550 

-0.05 

6SA7J 

250 

100 

0 

3.5 

8.5 

450 

-0.03 

6SA7  § 

250 

100 

-2 

3.5 

8.5 

450 

-0.03 

6K8 

250 

100 

-3 

2.5 

6.0 

350 

-0.08 

6L7 

250 

100 

-3 

2.4 

7.1 

375 

0.15 

*  This  is  the  capacitance  between  grid  and  cathode  with  all  voltages  applied  and  grid  biased  to 
plate-current  cutoff. 

t  This  is  the  increase  in  input  capacitance  as  the  grid  bias  is  varied  from  cutoff  to  the  plate  current 
indicated  in  this  table. 

$  Self  -excited. 

§  Separately  excited. 

The  reactive  component  of  input  admittance  is  essentially  capacitative  since  the  reac- 
tances due  to  the  lead  inductances  to  the  several  tube  elements  are  small  up  to  frequencies 
of  the  order  of  100  Me.  This  input  capacitance  consists,  in  the  typical  grid-controlled 
amplifiers,  of  the  static  paralleling  capacitances  between  the  signal  grid  and  all  grounded 
elements  plus  a  variable  capacitance  which  is  independent  of  frequency  but  changes  with 
tube  cathode  current  and  therefore  varies  with  the  tube  gain. 

The  variable  component  of  input  capacitance  may  be  broken  down  into  two  components. 
One  known  as  the  Miller  effect  is  due  to  the  grid-plate  or  feedback  capacitance,  Cgp.  With 
a  pure  resistance  output  load  in  the  plate  circuit  or  one  tuned  to  resonance  with  the  signal- 
input  frequency  the  added  input  capacitance  is 


(1 


(21) 


where  A  is  the  voltage  gain  of  the  stage.    In  a  triode  this  component  may  be  of  the  order 
of  1Q  to  100  times  as  large  as  the  other  input  capacitance  components.     Since  the  grid- 


VACUUM-TUBE   OPERATION 


4-19 


plate  capacitance  of  a  pentode  r-f  amplifier  tube  is  of  the  order  of  0.001  that  of  a  triode, 
this  Miller  effect  can  be  made  negligibly  small  with  a  properly  designed  pentode  amplifier. 
A  second  source  of  input-capacitance  variation  is  due  to  the  space  charge  in  the  cathode- 
grid  region.  Theory  predicts  that  the  capacitance  between  grid  and  cathode,  Cgk,  should 
increase  by  33  Vs  per  cent  when  space-charge-limited  current  flows  as  compared  to  no 
current  flow.  Measurements  show  that  there  is  an  increase  with  current  flow  but  not  an 
abrupt  one,  the  difference  being  due  in  all  probability  to  the  theory's  assumptions  of 
simple  geometry  and  zero  initial  velocity  of  emitted  electrons  not  being  fulfilled.  In 
Fig-  15  are  shown  graphically  the  results  of  measurements  of  input-capacitance  increases 


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FIG.  15.    Increase  in  Input  Capacitance  vs.  Grid-plate  Transconductance  for  a  Group  of  R-f  Pentode 

Amplifying  Tubes 

against  grid-plate  transconductance  for  a  group  of  typical  r-f  amplifier  pentodes.  These 
measurements  were  made  with  the  plate  grounded  to  radio  frequencies  at  100  Me.  Static 
voltages  are  as  indicated  in  the  previous  chart,  where  the  maximum  input-capacitance 
change  has  also  been  entered.  The  transconductance  was  varied  by  changing  the  grid  bias. 
COMPENSATING  FOR  INPUT-ADMITTANCE  CHANGES.  The  undesirable 
changes  of  input  admittance  as  the  gain  of  an  amplifier  tube  is  varied  may  be  made  reason- 
ably constant  by  the  introduction  of  negative  feedback  through  the  use  of  a  small  unby- 
passed  cathode  resistance,  rjt.  For  the  input  capacitance  to  have  the  same  value  at  full 
transconductance  as  it  has  at  cutoff,  the  value  of  rk  in  ohms  is  given  by 

ri  =  A^SputJL  (22) 


when  Cgk  is  the  capacitance  at  cutoff  between  grid  and  cathode,  ACinput  is  *ne  increase  in 
input  capacitance  from  cutoff  to  maximum  transconductance,  and  ggk  in  mhos  is  the  grid- 
to-cathode  maximum  transconductance  if  the  cathode  resistance  were  by-passed  with  a 
large  capacitor.  Both  fixed  and  maximum  variational  capacitance  values  are  given  in 
Table  1.  Note  that  this  compensation  holds  for  any  frequency. 


4-20 


ELECTRON  TUBES 


For  the  input  conductance  to  have  the  same  value  at  cutoff  as  it  has  at  full  gain,  the 
value  of  the  unbypassed  cathode  resistor,  r&,  in  ohms  is  given  by 


(23) 

where  ggk  in  micromhos  is  grid-cathode  transconductance,  Cgk  in  micromicrofarads  is 
grid-cathode  capacitance,  and  K  in  micromhos  per  megacycle  squared  is  the  frequency 
coefficient  of  input  loading.  This  compensation  is  also  independent  of  frequency. 
I  Generally  rk  will  not  have  the  same  value  for  both  capacitance  and  conductance  com- 
pensation, but  practically  they  are  of  the  same  order  of  magnitude  so  that  correcting  for 
one  will  usually  improve  the  other.  In  Fig.  16  are  shown  graphically  the  effects  on  two 
tube  types  of  adding  several  different  values  of  r& 


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FIG.  16.     Input-capacitance  Change  and  Input  Loading  vs.  Plate  Current  for  the  6AB7  and  6AC7 
Tubes.     (Courtesy  RCA  Review.) 

It  should  be  noted  that  the  unbypassed  cathode  resistance,  because  it  produces  de- 
generative amplification,  reduces  the  gain  to  1/(1  +  ggk?k)  of  its  value  when  r&  is  by-passed. 

NOISE  GENERATED  IN  VACUUM  TUBES.  The  vacuum  tube  is  a  noise  generator 
having  several  possible  sources.  One  important  type,  known  as  the  "shot  effect,"  stems 
from  the  fact  that  the  electron  current  consists  of  discrete  particles  which  leave  the  cathode 
in  a  random  fashion,  producing  fluctuation  currents  uniformly  distributed  over  all  fre- 
quencies. The  "flicker  effect"  is  a  low-frequency  phenomenon  caused  by  small  emitting 
areas  of  the  cathode  constantly  changing  their  emission  characteristics.  This  effect  is 
small  compared  to  the  shot  effect.  In  tubes  having  more  than  one  collector  element,  such 
as  the  screen  and  plate  of  a  pentode,  the  random  division  of  current  produces  uniform 
noise  currents  over  the  whole  frequency  spectrum  of  a  tube's  output  or  plate  current. 
Other  sources  are  positive-ion-emission  currents,  positive-ion  currents  produced  as  the 
result  of  gas  ionization,  and  secondary-electron  emission.  In  the  low-frequency  region 
may  also  be  found  microphonics  due  to  the  motion  of  the  tube  elements,  and  hum  result- 
ing from  the  use  of  an  a-c  power  source  for  heating  the  cathode.  Associated  with  the  input 
loading  and  therefore  present  only  at  high  frequencies  is  another  noise  source  which  may 
add  appreciable  noise  above,  say,  30  Me. 

The  thermal  agitation  or  shot  effect  for  a  temperature-limited  current,  /,  of  a  diode 
produces  a  mean-square  fluctuation  current  t2,  measured  in  a  frequency  band  width  A/,  as 
given  by  the  equation  _ 

#  =  3.18  X  l(rI9I  A/  (24) 

where  i  and  /  are  expressed  in  amperes  and  A/  is  in  cycles  per  second.  It  is  often  more 
convenient  to  express  the  tube-noise  generators  in  terms  of  an  equivalent  resistance  which 
at  room  temperature  produces  the  same  noise  as  the  hot-cathode  vacuum  tube.  The 
mean-square  thermal  agitated  fluctuation  current  for  any  short-circuited  pure  resistance  is 

(25) 


VACUUM-TUBE   OPERATION  4-21 

where  r  is  the  ohniic  value  of  the  resistance;  K  is  Boltzmann's  constant,  1.372  X  10 ~23 
joule  per  degree;  T  is  the  temperature  of  the  resistance  in  degrees  Kelvin;  and  i  is  in 
amperes.  Similarly  the  mean-square  fluctuation  voltage,  e2,  across  the  open-circuit 
terminals  of  a  pure  resistance  is 

?  =  4KTr  A/ 

where  e  is  now  given  in  volts.  If  room  temperature  is  assumed  to  be  290  deg  K  (81  deg 
Fahr)  the  above  expressions  become 


i2  =  ±^_d_±^ ^L  (26) 

e2  =  1.59  X  10  ^r  A/  (27) 

By  direct  substitution  the  equivalent-noise  resistance  of  the  diode  with  temperature- 
limited  current  flowing  is 

req  =  ~z  (28) 

Such  a  diode  is  conveniently  used  as  a  noise-signal  source  generator. 

With  the  diode  current  limited  by  space  charge,  and  provided  that  /  is  small  compared 
to  the  total  available  emission  current,  the  fluctuation  noise  currents  are  reduced  so  that 
the  equivalent  noise  resistance  for  the  diode  becomes 

req  =  ^  (29) 

301 

In  triodes  and  pentodes,  shot-effect  noise  is  present  as  in  the  diodes  and  may  be  repre- 
sented by  a  noise-equivalent  resistance  whose  thermal-agitation  noise  at  room  temperature 
is  equal  to  the  tube  noise  referred  to  the  control  grid  of  the  tube.  In  pentodes  the  random 
current  distributions  between  the  screen  grid  and  anode  produce  noise  usually  several 
times  greater  than  the  thermal  noise.  In  the  following  expressions  for  equivalent-noise 
resistances,  for  the  pentode,  the  first  term  in  the  parenthesis  is  due  to  shot  effect  and  the 
second  term  is  due  to  screen  current  fluctuation : 

For  triode  amplifiers, 

req  =  ~-  (30) 

For  pentode  amplifiers, 

J-  b  ~T~  •*  c2    \  S m  &m     / 

For  triode  mixers, 

req  =  |  (32) 

For  pentode  mixers, 

r*        '  Jf  +  2^\  (33) 

For  multigrid  converters  (with  inner-grid  or  outer-grid  injection), 


The  following  approximate  relationships  for  triode  and  pentode  mixers,  when  both  the 
oscillator  and  signal  frequencies  are  injected  in  the  grid  adjacent  to  the  cathode,  are  useful 
when  the  data  required  for  eqs,  (32)  and  (33)  are  not  available.  The  values  "as  amplifier" 
refer  to  conditions  at  the  peak  of  the  assumed  oscillator  cycle  which  is  usually  very  close  to 
zero  grid-bias. 

gc  (as  converter)  =  gm/4  (as  amplifier) 

/6  (as  converter)  =  /6/4  (as  amplifier) 

Ic2  (as  converter)  =  ZC2/4  (as  amplifier) 

req  (as  pentode  converter)  =  4req  (as  amplifier) 

T-gQ  (as  triode  converter)  —  6.5req  (as  amplifier) 

Conversion  from  noise-equivalent  resistance  to  noise-equivalent  rms  voltage  is  effected  by 
use  of  eq.  (27) : 

VP  =  1.3  X  10~10  Vreq  A/  (27a) 


4-22  ELECTRON  TUBES 

Conversion  from  noise-equivalent  resistance  to  noise-equivalent  rms  current  is  effected  by 
use  of  eq.  (26) :  

V?  -  1.3  X  1CT10  V—  (26a) 

^feq 

The  symbols  in  the  above  equations  have  conventional  significance  whose  definitions  are: 
req,  noise-equivalent  resistance,  ohms. 
gm,  grid-plate  transconductance,  mhos. 

gc,  conversion  transconductance  (frequency  converters  and  mixers),  mhos. 
J&,  average  plate  current,  amperes. 
JC2,  average  screen-grid  current,  amperes. 
Io,  average  cathode  current,  amperes. 
Ve2,  noise-equivalent  rms  voltage  for  band  width  A/,  volts. 

Vi2,  noise-equivalent  rms  current  for  band  width  A/,  amperes. 
A/,  effective  band  width,  cycles. 

In  Table  2  is  a  listing  of  representative  receiving-tube  types  showing  equivalent-noise 
values.  Several,  covering  a  large  range  of  equivalent-noise  resistance  values,  show  meas- 
ured values  all  in  good  agreement  with  values  computed  from  the  above  relationships. 

Positive-ion  noise  produced  by  collision  ionization  results  from  residual  gas  in  a  vacuum 
tube.  This  very  undesirable  fluctuation  noise  may  be  investigated  in  order  to  determine 
the  magnitude  of  grid  current  (positive-ion  current)  which  may  be  tolerated  from  analysis 
of  the  following  equation  for  the  equivalent-noise  resistance,  req  (gas) : 


req.  (gas)  =  (20^  +  A  -^  J  Iig     ohms 

\  Sin   / 


(35) 


where  r  —  grid-circuit  resonance  impedance,  ohms. 

Jo  =  cathode  current,  amperes. 

lig  =  positive-ion  current  to  grid,  amperes. 

gm  =  grid-plate  transconductance,  mhos. 

A  =  coefficient  of  the  order  of  40,000. 

The  term  20r2I^  represents  shot-effect  voltage  fluctuations  produced  by  the  gas  current 
flowing  in  the  grid  circuit.  It  may  be  increased  several  fold  by  induction  effects  associated 
with  the  ion  transit  time,  even  at  frequencies  of  a  few  megacycles.  The  second  term  is 
the  noise  generator  term  within  the  tube.  If,  for  an  example,  we  assume  r  as  IO6  ohms, 
gm  as  2000  X  10~6  mho,  Jo  as  10"2  amp,  and  a  grid  current  of  10~6  amp,  the  resulting 
req.  cgas)  is  250,000  ohms.  To  reduce  this  equivalent-noise  resistance  to  a  value  negligible 
compared  to  the  noise  produced  by  the  grid  circuit  resistance,  here  assumed  to  be  100,000 
ohms,  calls  for  a  reduction  of  the  gas  current  some  100  fold  or  to  10"2  microampere. 
Equation  (35)  is  applicable  to  triodes  or  pentodes. 

There  is  a  source  of  current  fluctuations  associated  with  the  component  of  tube  input- 
conductance  produced  by  electron  transit  time  effects,  a  source  which  becomes  important 
when  the  input  frequency  is  high  enough  to  make  the  transit-time  input  conductance 
relatively  large.  The  random  variations  in  space  current  will  induce  current  fluctuations 
in  the  control-grid  circuit,  giving  rise  to  grid-voltage  fluctuations  proportional  to  the  total 
input  impedance  (tube  and  circuit) .  These  induced  mean-square  grid-current  fluctuations 
may  be  expressed  thus: 


V«"f  (l-ljIXTuitf- 

Inserting  the  value  for  Boltzmann's  constant  K,  and  assuming  the  cathode  temperature, 
Tfc,  to  be  1000  deg  K,  the  above  expression  simplifies  to 

^2  =  7.5  X  lO"20^  A/    amperes  (36) 

The  noise-equivalent  rms  voltage  for  the  band  width  A/  appearing  at  the  tube  control-grid 
may  be  deduced  from  the  above  expression;  it  is 


2.75  X  IQ-iQ       f          volts  (37) 

S  +  gg 

when  gg  in  mhos  is  that  portion  of  the  tube  input  conductance  traceable  to  electronic  load- 
ing alone;  g  in  mhos  is  the  grid-circuit  resonance  conductance;  and  A/ in  cycles  is  the  effec- 
tive band  width  of  the  amplifier.  As  an  example,  assume  the  input  circuit  resonance 
impedance  to  be  20,000  ohms  so  that  its  reciprocal  g  is  50  X  10"*  mho;  let  gg  be  200  X 


VACUUM-TUBE   OPERATION 


4-23 


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4-24  ELECTRON  TUBES 

mho  and  assume  the  effective  band  width  to  be  5000  cycles.  The  rms  noise  voltage  appear- 
ing at  the  signal  grid  is  then  1.1  X  10  "^  volt,  certainly  not  negligible  for  a  high-gain 
amplifier! 

DISTORTION  INTRODUCED  BY  R-f  AMPLIFIER  TUBES.  Owing  to  the  inherent 
curvature  of  the  plate-current  vs.  signal-grid-voltage  characteristic  of  all  r-f  amplifier 
tubes,  there  are  present  in  the  tube  output  three  types  of  distortion  when  the  input  signal 
has  the  form  of  an  amplitude-modulated  carrier.  By  means  of  a  Taylor  expansion  series 
(see  also  Section  5,  articles  16-24),  the  plate  current  about  any  given  operating  point  may 
be  written 

r      ,        Mb 
^6  =  I,o  +  e,- 

In  this  expression  dib/dec  is  the  slope  or  transconductance,  gmQ,  of  the  plate-voltage  vs. 
grid-bias  curve,  and  dHb/dec2  and  S3ib/d3ec  are  the  second  and  third  derivatives  respectively 
of  this  same  curve.  The  quantity  eg  is  the  alternating  part  of  the  grid  voltage  or  the  signal 
input  voltage.  If  an  amplitude-modulated  carrier  signal  voltage  of  the  form 

eg  =  EI(\  -j-  ™>i  sin  pt)  sin  ccit 

is  inserted  in  the  above  series,  we  may  neglect  all  the  terms  having  harmonic  frequencies 
of  coi  :  if  it  is  assumed  that  there  is  a  tuned  circuit  in  the  output  of  the  tube  which  band- 
passes  only  those  frequencies  and  the  sideband  frequencies  associated  with  the  carrier 
frequency  OH.  Two  types  of  distortion  become  apparent  from  this  operation.  The  modu- 
lation factor,  mi,  is  changed,  and  the  modulation-frequency  component  has  amplitude 
distortion  which"  is  indicated  by  the  presence  of  harmonic  terms  of  p,  the  modulation 
frequency. 

The  modulation  factor  is  thereby  changed  by  the  ratio 


3    A 

8        /       gmo     J 

The  amplitudes  of  the  second  and  third  harmonics  producing  amplitude  distortion  of  the 
modulating  frequency  are  expressible  as:  (a)  Second  harmonic  distortion  (ratio  of  the 
amplitude  of  sin  2pt  to  the  amplitude  of  sin  pf)  . 

Ratio  of  2nd  harmonic  to  fundamental  —      —  mi  —  -  -  -     E\2  (40) 

l_16  gmQ     J 

(b)  Third  harmonic  distortion  (ratio  of  the  amplitude  of  sin  3pt  to  the  amplitude  of  sin  p£)  , 

Ratio  of  3rd  harmonic  to  fundamental  =     —  r  mi2  —  -  -  -     E-?  (41) 

L  32  gm0     J 

A  third  type  of  distortion  occurs  when  a  second,  and  usually  an  unwanted,  signal  modu- 
lates the  desired  signal.  This  is  known  as  "cross  modulation." 

If  there  is  substituted  in  the  above  series,  for  es,  eg  —  Ei(l  +  mi  sin  pt)  sin  u>it  + 
^2(1  +  W2  sin  q£)  sin  corf  where  the  subscripts  1  apply  to  the  desired  signal  and  the  sub- 
scripts 2  apply  to  the  undesired  signal,  then  the  cross-modulation  ratio  is 

Cross-modulation  ratio  =  |~i  —  dH*/de*^\  $?  (42) 

l_2  mi      gmQ     J 

Inspection  of  the  three  different  distortion  types  indicates  that  all  are  proportional  to  the 
square  of  the  signal  voltage  amplitude  and  that  all  also  are  proportional  to  the  third 
derivative  of  plate  current  to  grid  voltage  and  inversely  proportional  to  the  transconduct- 
ance. It  should  also  be  noted  that,  since  this  analysis  discards  all  higher  derivatives  than 
the  third,  reasonably  small  signal  voltages  are  implied. 

It  is  rather  tedious  to  obtain  sufficient  points  for  an  if,  vs.  e\  curve  to  determine  the  third 
derivatives  accurately  by  graphical  means.  A  more  practical  method  is  to  use  a  conven- 
tional gm  bridge  having  a  calibrated  and  variable  signal  grid  voltage.  The  procedure  is  to 
apply  a  small  signal,  of  the  order  of  0.01  to  0.1  volt,  to  the  signal  grid,  with  which  a  bal- 
ance is  established  which  gives  a  transconductance  reading  equal  to  gmQ,  The  bridge  is 
then  set  off-balance  by  a  predetermined  small  amount,  say  1  to  5  per  cent,  and  the  input 
signal  is  increased  until  the  bridge  is  again  in  balance.  The  required  change  in  gmQ  bal- 
ance may  be  either  positive  or  negative,  depending  on  the  curve  shape.  It  can  be  shown 
by  application  of  the  Taylor's  series  that 


8        gmQ 


VACUUM-TUBE   OPERATION 


4-25 


where  EQ  is  the  peak  value  of  the  large  signal  applied  to  the  grid.    Substitution  of  the  above 
into  the  previous  distortion  equations  results  in  the  following  expressions: 


Per  cent  change  in  modulation  =  2(1 mi2  )  — —  X  100 

\         8        /  gmQ 

Per  cent  2nd  harmonic  distortion  —  ~mi  — ^  X  100 

2  gmQ 


Per  cent  3rd  harmonic  distortion  =  -  mi2  — — 

4         gmo 

Per  cent  cross  modulation  =  4 —  X  100 


100 


If  it  is  assumed  that  mi  —  I,  that  m%/mi  —  1,  and  that  the  transconductance  is  changed 
1  per  cent,  the  above  may  be  expressed  to  show  the  relative  magnitudes  of  the  various 
distortions,  giving 

Modulation  change  =  1.25% 

2nd  harmonic  distortion  =  1.5% 

3rd  harmonic  distortion  =  0.25% 

Cross  modulation  =  4% 

In  Fig.  17  are  plotted  the  signal  voltage  necessary  to  change  the  gm  by  3  per  cent  vs.  the 


Signal,  in  volts  rms,  to  give  a 
3  per  cent  change  in  transconductance 

OO  f-»  K>  UJ  4^>  Ui  C 

1 

\ 

6K7    maximum  transconductance  = 
6AB7         '                      " 

=  1450  jumhos 
=  5000  jumhos 

1 

\ 

1 

\ 

1 

6K7H 

I 

\ 

1 

\ 

F 

V 

\ 

/ 

1 

\ 

** 

•s, 

y 

1 

\ 

^ 

**<*» 

N 

s 

\ 

\ 

6AB7/ 

^ 

\ 

s 

V. 

/ 

\ 

\ 

***** 

+  <^ 

"\ 

/ 

.1                                         1.0                                          10                                         1C 

Transconductance  in  percent  of  maximum  transconductance 

FIG.  17.    Comparison  of  Signal  Voltages  Applied  to  the  Remote  and  Semi-remote  Cutoff  Tube  Types 
6K7  and  6AB7  Respectively  to  Produce  the  Same  Amount  of  Distortion 

transconductance  in  terms  of  the  maximum  transconductance.    For  modulation  factors 
of  unity,  this  corresponds  to  saying  that  the  signal  voltages  indicated  will  produce 

Modulation  change  —  3.75% 

2nd  harmonic  distortion  =  4.5% 

3rd  harmonic  distortion  =  0.75% 

Cross  modulation  =12% 

The  one  curve  is  for  a  "remote  cutoff"  tube,  6K7,  which  requires  approximately  45  volts 
bias  to  reduce  its  maximum  transconductance  of  1500  jLonhos  to  about  10™3  this  value. 


4-26 


ELECTRON  TUBES 


The  second  curve  is  for  a  "semi-remote  cutoff"  tube,  6AB7,  which  requires  about  22  volts 
bias  to  reduce  its  maximum  transconductance  from  5000  to  10~3  this  value.  It  is  apparent 
that  the  6K7  can  handle  a  much  larger  signal  than  the  6AB7.  The  6AB7  tube  is  limited 
to  lower  signal  values  because  of  its  higher  transconductance  value,  and  if  it  were  designed 
to  have  a  similar  cutoff  as  the  6K7  it  would  draw  exceedingly  large  plate  currents  when 
the  bias  was  set  low  for  maximum  gain.  There  is  thus  a  practical  limitation  governing 

the  maximum  transconductance  and 
maximum  signal-handling  capacity 
that  can  be  designed  into  a  tube. 

CHANGING  OPERATING  CON- 
DITIONS. Operating  Voltages.  It 
is  sometimes  necessary  to  change  the 
operating  potentials  of  an  amplifier 
or  a  power  output  triode  or  pentode 
tube  from  the  published  typical  op- 
erating conditions.  By  means  of  the 
conversion  factor  chart,  Fig.  18,  it  is 
possible  to  determine  the  new  volt- 
ages, currents,  transconductance, 
plate  and  load  resistances,  and  power 
output.  The  curves  are  based  on 
the  fact  that,  if  all  applied  voltages 
(except  heater  voltage)  are  changed 
by  a  factor  n,  the  resulting  currents 
will  all  be  changed  by  ns/2,  the  gm  by 
n1/2,  and  the  plate  resistance  by  n5jz. 
The  accuracy  of  the  chart  is  reason- 
ably good  for  small  changes,  but  for 
large  voltage  changes  exceeding  2.5 
to  1  the  chart  is  unsuitable.  As  an 
example,  assume  a  pentode  with  the 
following  typical  ratings : 

Plate  voltage 250  volts 

Screen  voltage 250  volts 

Grid  bias ? - 18  volts 

Plate  current 32  ma 

Screen  current 5.5  ma  - 

Plate  resistance 70,000  ohms 

Transconductance 2,300  Aimhos 

Load  resistance 7,600  ohms 

Power  output 3.4  -watts 

j 

3.0 


1U 

8 
6 

4 
3 

u? 

E 
2  1.0 

0 

£0.8 

.2 
gO.6 

c 
o 
O 
0.4 

0.3 
0.2 

n.i 

/ 

i 

1 

I 

j 

/ 

I 

/ 

i 

y 

^ 

\ 

"V, 

^ 

i 

^ 

^t 

n 

^^^^ 

*s 

^ 

/ 

I 

"^^ 

s* 

// 

"\^ 

s* 

/ 

/ 

\ 

/ 

1 

/ 

1 

/ 

/ 

/ 

F, 

applies  to  plate,  screen, 
and  contro   grid  voltage.              — 
applies  to  plate  and  screen 
current, 
applies  to  power  output, 
applies  to  plate  resistance           ~~ 
and  load  resistance, 
applies  to  mutual 
conductance. 

/ 

/ 

Fp 

/ 

/ 

* 

jm 

0.4         0.6    0.8  1.0        1.5     2.0 
Voltage  Conversion  Factor,  Fe 

FIG.  18.     Conversion  Factor  for  Triode  and  Pentode 


It  is  desired  to  determine  the  opera- 
tion characteristics  for  a  plate  volt- 
age of  100  volts.  All  voltages  will 
have  to  be  changed  in  the  ratio  of 
100  to  250  (e.g.,  Fe,  the  voltage  conversion  factor,  is  0.4).  From  the  chart  the  new  to  the 
old  ratios  are  picked  off  so  that  Ft  is  0.25,  Fp  is  0.1,  Fgm  is  0.63,  and  Fr  is  1.6.  The  new 
conditions  will  then  be 

Plate  voltage 100  volts 

Screen  voltage 250  X  0.4       =100  volts 

Grid  bias - 18  X  0.4     =  -7.2  volts 

Plate  current 32  X  0.25       =  8  ma 

Screen  current 5.5  X  0.25       =  1.4  ma 

Plate  resistance 70,000  X  1.6  =  112,000  ohm 

Transconductance 2,300  X  0.63  =  1,450  /tmhos 

Load  resistance 7,600  X  1.6    »  12,000  ohms 

Power  output 3.4  X  0.1         =  0.34  watt 

It  should  be  noted  that  this  chart  cannot  be  used  if  only  one  voltage  is  varied. 

Changes  in  Heater  or  Filament  Voltage.  Changing  the  heater  or  filament  voltage  of  a 
tube  will  increase  or  decrease  the  temperature,  current,  and  power  input  of  the  emitting 
cathode.  The  curves  of  Fig.  19  show  these  relationships,  which  were  based  on  taking 
average  values  of  radiation  coefficients  and  resistivity  of  tungsten,  molybdenum,  tantalum, 
and  nickel  covering  temperature  ranges  of  1000  to  2800  deg.  K.  With  a  maximum  fila- 
ment voltage  variation  of  ±25  per  cent,  engineering  accuracy  holds  for  vacuum  and 


VACUUM-TUBE  OPERATION 


4-27 


gas-filled  tubes,  the  greatest  deviation  between  observed  and  predicted  currents  and  power 
input  being  only  4  per  cent. 

As  an  example  assume  that  a  6.3  volt  0.3  amp  heater  is  running  with  a  cathode  tempera- 
ture of  1050  deg  K  and  a  heater  temperature  of  1400  deg  K  and  that  the  heater  voltage  is 
decreased  to  5.5  volts  or  87.3  per  cent  of  nominal.  From  Fig.  19  we  find  that  the  resulting 
power  input  is  80  per  cent  of  1.89  watts,  or  1.51  watts.  The  new  filament  current  is  re- 
duced to  92  per  cent  of  0.3  amp,  or  0.276  amp,  and  the  cathode  and  heater  temperature 
are  reduced  to  95.8  per  cent,  or  1005 
and  1340  deg  K,  respectively. 

Maximum  Allowable  Grid  Resist- 
ance. Common  practice  is  to  apply 
the  d-c  bias  to  the  negative  control 
grid  of  an  amplifier  through  a  series 
grid  resistor.  It  is  desirable  to  make 
this  resistor  as  high  valued  as  possible 
since  it  shunts  the  signal  source  and 
therefore  absorbs  power  and  may  have 
detrimental  results  on  the  frequency 
response  and  loading  of  the  signal 
source.  Its  limiting  value  is  usually  of 
the  order  of  0.1  to  10  megohms  and  is 
established  by  the  radio-tube  manu- 
facturer on  the  basis  of  life  tests  and 
maximum  expected  grid  current.  Grid 
current  flowing  through  the  grid  resis- 
tor decreases  the  grid  bias  by  the  IR 
drop  through  the  resistor  and  may 
cause  the  tube  to  "run  away"  since 
the  resulting  increase  in  plate  current 
will  produce  excessive  plate  dissipa- 
tion. By  the  addition  of  a  cathode 
self-bias  resistor,  or  the  use  of  a  series 
dropping  resistor  in  the  screen  lead  of 
a  pentode,  or  the  use  of  a  d-c  load  re- 
sistor in  the  plate  lead  of  a  triode, 
there  results  some  d-c  degeneration, 
thereby  making  it  possible  to  increase 
the  value  of  the  grid  resistor  above 
that  indicated  as  a  maximum.  Also, 
if  the  tube  is  operated  at  a  reduced 
value  of  transconductance,  the  maxi- 
mum allowable  grid  resistance  value 
may  be  increased. 

From  the  following  equation  it  is 
possible  to  determine  the  maximum  allowable  value  of  grid  resistance,  rgi,  for  any  new  set 
of  operating  conditions  differing  from  those  published. 


Per  cent  of  basic  watts,  current,  temperature 

^OTO>->JOOU)Ol-'  10  W  £»  01  O>  v 

ooo  ooooo  oooooc 

/ 
f 

/ 

f 
/ 

/ 
*~~/ 

/ 

/ 

/ 

&**~ 

/ 

/ 

^ 
^ 

^ 

^ 

*^e 

tf& 

*s 

*7 

""cfe^ 

/" 

/ 

/ 

/' 

/ 

#/ 

w 

/ 

/ 

f 

/ 
^/ 

50       70        80        90       100      110      120      130      14 

Per  cent  of  basic-voltage 

Chart  Giving  Wattage,  Current,  and  Temper- 
,  Heater,  or  Cathode  at  Operating 


FIG.  19. 

ature  of  a  Filament,  '. .  ,   _..    . ,  _ 

Voltages  up  to  25  Per  Cent  above  or  below  Basic  Voltage, 

with  Sufficient  Accuracy  for  Most  Engineering  Purposes. 

Accuracy  drops  in  dotted  regions. 


(44) 


where,  for  a  triode, 

A7VAIffl  is  the  ratio  of  plate  current  change  per  unit  change  in  grid  current; 

gk  is  the  grid-plate  transconductance  in  mhos; 

r  is  the  d-c  plate  load  resistance; 

rp  is  the  internal  tube  plate  resistance; 

rjc  is  the  cathode  self-bias  resistance  in  ohms; 

p  is  the  triode  mu; 
and  where,  for  a  pentode, 

A7fc/AIgl  is  the  ratio  of  cathode  current  change  per  unit  change  in  grid  current; 

gjc  is  the  signal-grid-to-cathode  transconductance  in  mhos  (this  may  be  determined  by 
multiplying  the  signal-grid-to-plate  transconductance  by  the  factor  (Ip  ~h  Ic^/Ip)'* 

r  is  the  screen  dropping  resistance; 

TP  is  the  internal  tube  screen  resistance; 

Tk  is  the  cathode  self-bias  resistance,  in  ohms; 

{j,  is  the  triode  connected  amplification  factor,  i.e.,  the  mu  from  control  grid  to  screen 
grid  of  the  pentode. 


4-28 


ELECTRON  TUBES 


CUtho-d-e  Mis    |        1.4 


2.0     1        2.5-5.0        | 


6.3 


|        12.6-117 


Kinescopes 


Projec- 
tion 

magnetic  deflection 

5TP4 

Directly 

Viewed 

magaetie  deflect  ion 

9AP4 
12AP4 

7DP4 
10BP4 

electrostatic 
dfrlectioon 

7JP4 

Rectifiers  (for  rectifiers  with  amplifier  units,  see  Power  amplifiers) 


Half- 
wave 

Y3C1LU.K1 

IB3-GT/ 
8016* 

1-v 
81  f 

12Z3F    35W4    1 
35  Y4    35Z4-GT 
35Z3L35Z5-GTJ 
45Z3    45Z5-GT 
11  723 

Full- 
wave 

vacuum. 

5T4,  5W4 
r5U4-G,5X4-G] 
I        5Z3        j 
T  5Y3-GT  1 
5Y4-G 
L      80      J 
5Z4 
[5V4-G,  83-v] 

f      6X4,  6X5      I 
L6X5-GT,  84/6Z4J 
6ZY5-G 
7Y4 
7Z4 

raereiuy-T^por 

82       83 

gas 

Cold-Cathode  Types:  OZ4,  OZ4-G 

Dcrabler 

vacuum 

f    25Z5    I 
25Z6 
|_25Z6-GTJ 
50Y6-GT 
117Z6-GT 

Diode  -detecctow  C/or  diode  detectors  with  amplifier  units,  see  Voltage  amplifiers  and  also  Power  amplifiers) 


One  diode 


1A3 


I 


6AL5  [6H6,  6H6-GTJ  7A6   |     12H6    12AL5 


ler  amplifiers  with  and  without  rectifiers,  diode  detectors,  and  voltage  amplifiers 


Triodes 

low-ma 

snngle  innit 

31 
49 

2A3     45 
46     71-A 

6B4-G    10f    6A5    50  1 

b  igh-o"U 

snngle  iBnit 

6AC5-GT 

tinn  unit 

1G6-GT 

T1J6-G1 
L    19    J 

53 

[6A6,  6N71    A77  r    70 
L6N7-GTJ    6Z7'G    79 

direot-eoirpled 

arrangement 

6B5 

Beam 
Tubes 

single  unit 

f  1Q5-GT  1 
L3Q5-GT  tl 
1T5-GT 
3LF4I 

6BG6-G          T  6AQ5   ~| 

[6L61         L6V6-GTJ 
L6L6-GJ 

6Y6-G      7A5      7C5 

14A5  r   25L6   1 
35A5  L25L6-GTJ 

[35B5  35L6-GT] 
50A5 

[50B5,  50L6-GT] 

witk  recfciffier 

32L7-GT 
70L7-GT 
ri17L7/M7-GTl 
L    117P7-GT    J 
117N7-GT 

Pen- 
todes 

single  unit 

1A5-GT 
1C5-GT 
1LA4,  1LB4 

[1S4,  3S4|] 
[3Q4t,3V4JI 

f  1F4  ] 
L1F5-GJ 
1G5-G 
1J5-G 
33 

2A5 
47 
59 

6A4/LA  [6AK6,  6G6-G]  6AG7 

[6F6,  6F6-G,  6F6-GT,  42] 
[6K6-GT,  41] 

7B5       .38        89 

[25A6] 
L  43  J 

with  nwdruiHEni 
triode 

6AD7-G 

with,  diode  ani 
triode 

1D8-GT 

with.  recfciSer 

12A7 

twk  unit 

1E7-G 

*  Cathode  volts,  1.25.       f  Cathode  volts,  7.5. 
PIG.  20.    Receiving  Tube  Classi- 


VACUUM-TUBE   OPERATION 


4-29 


Cathode  Volts     |        1 .4 


2.0       |2.5-5.0| 


6.3 


12.6-117 


Converters  &  mixers  (for  other  types  used  as  mixers,  see  Voltage  amplifiers) 


Con- 
vert- 
ers 

pentagrid 

1A7-GT 
1LA6 
1LC6 
1R5 

[  1C6  ] 
11C7-GJ 
r  1A6  1 
11D7-GJ 

2A7 

T      6A7,  6A8      1  [6BE6,  6SA7] 
6A8-G,  6A8-GT     L  6SA7-GT  J 
L       6DS-G        J 
6SB7-Y        7B8        7Q7 

12A8-GT 
[12BE6,  12SA71 
L    12SA7-GT    j 
14B8      14Q7 

triode-hexode 

[6K8,  6K8-G] 

I2K8 

triode-heptode 

6J8-G         7J7         7S7 

14J7 

octode 

7A8 

Mixers 

pentagrid 

[6L7,  6L7-G] 

Electron-ray  tubes 


Single 

with  remote-cutoff 
triode 

6AB5/6N5         6U5/6G5 

with  sharp-cutoff 
triode 

2E5 

6E5 

Twin 

without  triode 

6AF6-G 

Voltage  amplifiers  with  and  without  Diode  Detectors;  Triode,  tetrode,  and  pentode  detectors;  oscillators 


Triodes 

medium- 
mu 

single  unit 

1G4-GT 
1LE3,  26  § 

[1H4-G1 
L    30    J 

27 
56 

6C4  [6C5,  6C5-GT]  [6P5-GT,  76] 
[6J5,  6J5-GT]  6L5-G,  7A4,  37 

12J5-GT 
14A4 

with  r-f 
pentode 

6F7 

with  power 
pentode 

6AD7-G 

with 
pentode 
and 
diode 

1D8-GT 
3A8-GT 

with,  two 
diodes 

T1B5/25S] 
L  1H6-G  J 

55 

[    6R7.6R7-GT     1  7FA    R, 
L6BF6,  6SR7,  6ST7J  /Jit)    w 

f    I2SR7     I 
L12SR7-GT  J 

14E6 

twin  unit 

6C8-G  [6F8-G,  6SN7-GT]  6J6 
7N7    7F8    12AU7 

12AH7-GT 
12AU7  12SN7-GT 
HN7 

high-mu 

single  unit 

f  6F5,  6F5-GT  1    ,Trr  PT  «,, 
L6SF5,  6SF5-GTJ    6K5"GT  /B4 

T12F5-GT] 
L   12SF5  j 

with  diode 

1H5-GT 
1LH4 

with  two 
diodes 

2A6 

F6SQ7,  6SQ7-GT]  r  6AT6,  6AQ6  ' 
L     6B6-G,  75     J      6Q7,  6Q7-G 
6T7-G,  7B6,  7C6  !_6SZ7,  6Q7-GT. 

[12AT6,  I2Q7-GTJ 
14B6 
[12SQ7r  12SQ7-GT] 

with  three 
diodes 

6S8-GT 

twin  unit 

6SC7    6SL7-GT    7F7    12AX7 

12SC7    I2AX7 
12SL7-GT  14F7 

Tet- 
rodes 

remote  cutoff 

35 

sharp  cutoff 

32 

24-A 

36 

Pen- 
todes 

remote 
cutoff 

single  unit 

IT4 
1P5-GT 

34 
riD5-GPl 
L  1A4-P  J 

58 

F6K7,  6K7-G1  P  6D6  ][6BA6,6SG7] 
L6K7-GT,  78J  [6U7-GJ      6BJ6 
6AB7/1853    [  6S7  1    7A7,  7B7 
r   6SK7    ]   L6S7-GJ       7H7 
L6SK7-GTJ     6SS7          39/44 

[12BA6,  12SG7] 
f    12SK7    I 
L12SK7-GTJ 
12K7-GT,  14H7 
14A7/12B7 

with  triode 

6F7 

with  diode 

6SF7 

12SF7 

with  two 
diodes 

2B7 

[dBS,  6B8-G]7137    7R7 

12C8,  14R7 

sharp 
cutoff 

single  unit 

1LC5,  1LN5 
1L4,  1U4 
1N5-GT 

PE5-GP1 
L  1B4-P  J 
15 

57 

F6J7,  6J7-G,  6J7-GT]  f    6SJ7    1 
L  606,  6W7-G,  77   J  L6SJ7-GTJ 
r6ATI61   6AC7/1852       6AG5 
L6SH7J    7G7/1232          7C7 
7L7            7V7              7W7 

[I2AU6,  12SH7] 
12AW6 
f    12SJ7    ] 
U2SJ7-GTJ 
12J7-GT  14C7 

with  diode 

1LD5 
[1S5,  1U5] 

with  two 
diodes 

r  1F6  i 
L1F7-GJ 

t  Filament  arranged  for  either  1.4-  or  2.8-volt  operation, 
fication  Chart.   (Courtesy  RCA.) 


§  Cathode  volts,  1.5. 


4-30  ELECTRON  TUBES 

All  the  above  factors  can  usually  be  obtained  from  published  tube  characteristics  or  are 
measurable  for  a  given  tube  type  with  the  exception  of  the  ratio  A/fc/AIgl.  This  ratio 
indicates  that  for  any  given  amount  of  grid  current  flow  there  is  a  definite  increase  in 
cathode  current  flow.  If  it  were  possible  to  manufacture  vacuum  tubes  so  that  with  a 
negative  grid  absolutely  no  grid  current  flowed,  the  series-grid  resistance  could  be  infinite 
in  value.  However,  for  practical  tubes,  grid  currents  of  the  order  of  a  microampere  may 
exist  due  to  residual  gas  ionization,  thermal  and  photoelectric  emission,  and  leakage  cur- 
rents. This  ratio,  AIjb/AI^,  for  a  given  tube  may  be  determined  from  the  published  max- 
imum allowable  grid  resistance  by  substituting  it  in  the  above  equation,  and  once  de- 
termined it  may  be  used  to  determine  a  new  value  of  maximum  resistance  for  a  new 
operating  condition. 

Example.    A  pentode  with  fixed  bias  and  fixed  screen  voltage  has  indicated  a  maximum  allowable 
grid  resistance  of  0.2  megohm.    What  is  the  maximum  grid  resistance  with  full  self-bias  and  a  series 
dropping  resistance  from  a  300-volt  supply?    What  is  the  maximum  grid  resistance  if  the  bias  is  in- 
creased so  that  the  cathode  current  is  reduced  to  one-tenth  its  maximum  value? 
For  rgi  —  0.2  megohm, 

Egl  «=*  —  2  1/2  volts 
EC2  —  100  volts 
ECZ  —  0  volts 
Eb  =  250  volts 


M12  =  25 

r2  —  25,000  ohms  (internal  screen  resistance) 
z"c2  =  2.5  ma 
ip  —  10  ma 

The  cathode  transconductance,  gift,  is  determined  by, 

„  =  ^  «L±^> 

Then  n  =  —  -  (  —  )  ,  since  there  are  no  self-bias  or  screen  dropping  resistors  used.    Then 
Al£l  \gk/ 

^L  =  rigk  =  (0.2  X  106)(6250  X  10~6)  =  1250  (46) 

Al*i 

For  the  new  conditions  the  self-bias  and  series  dropping  screen  resistors  are 
rk  =  2.5/(12.5  X  10~3  amp)  =  200  ohms 

r£2  =  (300  —  100)/(2.5  X  10~3  amp)  =  160,000  ohms  series  dropping  screen  resistor 
The  new  maximum  grid  resistance  is 

^-^[&0  +  aJ^Mo)^(^g)] 

=  632,000  ohms 

At  one-tenth  normal  cathode  current  the  new  bias  is  computed  to  be  approximately  —3.7  volts,  assum- 
ing that  the  tube  obeys  the  three-halves  power  of  effective  voltage  law.    The  new  values  are  then 

rk  »  3.7/(1.25  X  10~3  amp)  =  2950  ohms 
ri2  =  160,000  X  10  =  1.6  megohms  series  screen  resistor 
Tg2  =  25,000  (10)^  =  53,000  internal  screen  resistance 

gk  =  6250  X  10~6/10^  =  2910  X  10~6  mho 
From  the  above  the  new  maximum  grid  resistance  is 

•*"*»  [^(^leT^)^850^^)] 

«=  4.7  megohms 

RECEIVING  TUBE  CLASSIFICATION  CHART.  Figure  20  classifies  the  commonly 
used  receiving  tubes  according  to  their  functions  and  their  cathode  voltages.  It  is  arranged 
to  permit  quick  determination  by  the  tube  user  of  the  type  designations  of  tubes  applicable 
to  specific  design  requirements.  Types  having  similar  characteristics  and  in  the  same 
cathode-voltage  groups  are  bracketed. 


TYPICAL  VACUUM-TUBE   CHARACTERISTIC  CURVES      4-31 


6.  TYPICAL  VACUUM-TUBE  CHARACTERISTIC  CURVES 

The  following  vacuum-tube  characteristic  curves  were  selected  as  representative  of  the 
triode  (type  10),  the  tetrode  (type  865),  and  the  suppressor-grid  pentode  with  normal  (type 
57)  and  with  remote-cut-off  control  grid  (type  58) .  The  data  were  furnished  by  the  RCA 
Radiotron  Division,  RCA  Mfg.  Co.,  Inc. 


CD      8      °. 


iffi 


.2     T3 
i      O 


4-32 


ELECTRON   TUBES 


-  /u 

.  L._ 

pe-  10 
je  Transfe 
cterist  cs 
5  Volts  D.C. 

TV 

j 

/ 

/ 

Pf-iar- 

t 

/ 

E/=7. 

^ 

7 

I 

50 

r                 / 

1 

- 

I 

7 

j(                 J 

--    Z 

7                  7 

7 

j                   7 

I 

7                  7 

r 

f 

/ 

40   « 

_j          y 

i 

/               /       n 

Q. 

j 

E 

.55 

2            j 

^ 

-      I  -_^  , 

t 

"  "1 

r              r 

/ 

y 



-CO 

/              2 

7 

J2 

ijj  I 

-/ 

r 

j              l 

J 

(f 

~l 

~^3.  

-J 

-^  r  — 

£           v^ 

T_ 

-  ..  .  L 

Si 

^ 

^ 

H                           f 

,                1 

<r 

;7                7 

7                   ^ 

1 

/ 

f-                r 

/                   / 

' 

c?  / 

J_JL 

) 

*£*/ 

f 

I 

20 

/_j 

"V7 

_  7               IA_ 

7 

f              c 

S                 A 

S               r 

y 

/ 

i 

7                     , 

^ 

7 

f 

/ 

; 

_              _i 

JL                           JL 

i  i 

/ 

r  "" 

{__ 

7 

/ 

V 

<?/- 

10 

7 

7                           ^ 

i 

/ 

y 

7 

7 

2                           2 

r 

^ 

7 

Q 

^ 

T 

/ 

?            •"            ? 

2 

_j  —  L^  !  

—  r-  1  

r- 

7 

L  ^t.  

/  H7~ 

-/- 

/               I 

«r-                  ^- 

_     ^  ' 

E             ^ 

/ 

7    (    ~y" 

_,  _ 

X  -7  

rt 

SO                         T-8O                          —60                          —4O 
Grid  Volts 

^20                               0" 

FIG.  22.     Typical  Triode  (Type  10) 


TYPICAL    VACUUM-TUBE    CHARACTERISTIC    CURVES       4-33 


2000         400 


20  40 

Plate  MiJIiamperes 

FIG.  23.     Typical  Triode  (Type  10) 


4-34 


ELECTRON  TUBES 


lype 
Average  Plate 
Characteristics 


400 
Elate  Volts 


.800 


FIG.  24.     Typical  Tetrode  Characteristics  (Type  865) 


TYPICAL  VACUUM-TUBE   CHARACTERISTIC  CURVES     4-35 


Type  -  865 

Average 

Characteristics 

E/»7.5  Volts  D..C. 

Screen  Vo!ts=125  D.C. 


22 


20 


18 


1-6 


£ 

14  I 


12 


-60 


0 


-50  -40  -30  -20  -10  0  10 

Control  Grid  Volts 

FIG.  25.    Typical  Tetrode  Characteristics  (Type  865) 


4-36 


ELECTRON   TUBES 


Ttoe  -  865 
Average 
Characteristic 
Ef=7.5  Volls  D.C 
Plate  Vo  ts=  50O  C 

^ 

/ 

-i 

/" 

-T^l 

"3  

/ 

j 

f  ~ 

-25 

s 

~~^i_ 

f- 

7 

t 

Q 

/ 

-$ 

f- 

oc 

JL 

d. 

J 

/i 

y- 

-/- 

(_ 

J  — 

*o 

^ 

5f 

^ 

-J-- 

n  

I 

T 

y 

7 

/ 

( 

.S 

/ 

t- 

7- 

1} 

-T| 

-^ 

y 

-  15 

/ 

/ 

/ 

^ 

; 

/ 

/ 

^     • 

/ 

/ 

/ 

/ 

/ 

-^  

^ 

/ 

/ 

7 

/ 

-  10 

~7~ 

-^L 

7 

/- 

>    - 

/ 

/ 

7 

7 

/ 

Iz 

X 

-  «? 

-f*- 

__x 

^  — 

y 

— 

x 

/ 

r 

5 

-?^ 

2 

-f 

& 

o 

Tr 

Si 

S 

*fa 

re 

i 

0 

p> 

I 

t  _ 
=  — 

M 

e£ 
& 

=  = 
••*! 

] 

W 

5C 

FIG 

£—  r 

1 

) 
.    2 

6. 

?* 
^ 

= 

5= 

S 

T5 

===  = 

=  !=•=: 

^ 

ST 

^ 

= 

H 

L30- 

ISM—  ft    i    i    i    i    iti..  i    i    i    i    i    i    i    i    i    i    i    i    i    i    i    i    i    i    i    i 
100                            150                            200 
Screen  Grid  Volts 

Apical  Tetrode  Characteristics  (Type  865) 

250 

TYPICAL    VACUUM-TUBE    CHARACTERISTIC    CURVES       4-37 


:  i> 


4-38 


ELECTRON  TUBES 


Type  -  57 

Average 

Characteristics 

E/=2.5  Volts 
Screen  Volts  =  100 
Suppressor  Volts  =0 
Plate  Volts  =250 


20'00 


16OO 


1400 


1200  § 


1000 


40O 


200 


_,7  —a  -5  -4,  -3  -.2.  -1 

CAD.tf.oJ  GxLd  VStts 

FIG.  28.     Typical  Pentode  Characteristics  (Sharp  Cutoff  Type  57) 


0 


TYPICAL    VACUUM-TUBE    CHARACTERISTIC    CURVES       4-39 


2000 


1800 


1600 


1400 


I 


600 


400 


200 


Average  Chi-atracterf sties 
S/-2.5  Volts 
Screen  Voffs*  100 
Suppressor  Volts=0 
Plate  volts=250 


800 


FIG.  29.     Typical  Pentode  Characteristics  (Remote  Cutoff  Type  58) 


4-40  ELECTRON  TUBES 


BIBLIOGRAPHY 

Llewellyn,  !F.  B.t  Electron-Inertia  Effects,  Cambridge  University  Press,  Cambridge  (1943). 

Ferris,  W.  R.,  Input  Resistance  of  Vacuum  Tubes  as  Ultra-high-frequency  Amplifiers,  Proc.  I.R.E., 

Vol.  24,  82  (January  1936). 
North,  D.  OM  Analysis  of  the  Effects  of  Space  Charge  on  Grid  Impedance,  Proc.  I.R.E.,  Vol.  24,  108 

(January  1936). 
Strutt,  M.  J.,  and  A.  Van  der  Ziel,  The  Causes  for  the  Increase  of  Admittances  of  Modern  High- 

frequency  Amplifier  Tubes  on  Short  Waves,  Proc.  I.R.E.,  Vol.  26,  1011  (August  1938). 
Rothe,  H.,  and  W.  Kleen,  Grundlagen  and  Kennlinien  der  Electronenrdhren,  Akademisehe  Verlagsgesell- 

schaft,  Leipzig  (1943).    Also  by  the  same  authors  and  publisher,  Elecktronenrohren  als  Anfangsstufen- 

Verstarkes  (1940). 
Langmuir,  L,  and  K.  T.  Compton,  Electrical  Discharges  in  Gases,  Part  II.    Fundamental  Phenomena 

in  Electrical  Discharges.  Rev.  Modern  Phys.,  Vol.  3,  191  (April  1931).  This  is  an  excellent  paper. 
Herold,  E.  W.,  The  Operation  of  Frequency  Converters  and  Mixers  for  Superheterodyne  Reception, 

Proc.  I.R.E.,  Vol.  30,  84  (February  1942). 
Thompson,  B.  J.,  D.  0.  North,  and  W.  A.  Harris,  Fluctuations  in  Space-charge-limited  Currents  at 

Moderately  High  Frequencies,  R.C.A.  Rev.,  Vol.  4,  269-285  (January);  Vol.  4,  441-472  (April); 

Vol.  5,  106-124  (July);  Vol.  5,  240-260  (October  1940);  Vol.  5,  371-388  (January  1941). 
North,  D.  O.r  and  W.  R.  Ferris,  Fluctuations  Induced  in  Vacuum-tube  Grids  at  High  Frequencies, 

Proc.  I.R.E.,  Vol.  29,  49-50  (February  1941). 
Herold,  E.  W.,  Superheterodyne  Converter  System  Considerations  in  Television  Receivers,  R.C.A.  Rev., 

Vol.  4,  324-337  (January  1940).    This  article  gives  general  expressions  for  conversion  trans  conduct- 

ance and  equivalent  noise. 

Haller,  C.  E.,  Filament  and  Heater  Characteristics,  Electronics,  July  1944,  pp.  126-130. 
Ballantine,  S.,  and  H.  A.  Snow,  Reduction  of  Distortion  and  Cross-talk  in  Radio  Receivers  by  Means 

of  Variable-mu  Tetrodes,  Proc.  I.R.E.,  Vol.  18,  12,  pp.  2102-2107  (December  1930). 
Herold,  E.  W.,  R-f  Distortion  or  Cross-modulation  of  Pentode  Amplifier  Tubes.     Electronics,  April 

1940,  pp.  82-88. 

MAGNETRONS 

By  W.  B.  Hebenstreit 

In  this  section  only  those  magnetrons  of  circular  cylinder  geometry  will  be  considered. 
A  cylindrical  anode  is  coaxial  to  an  emitting  cathode,  and  the  elements  are  mounted  in  a 
vacuum  envelope.  A  static  magnetic  field  is  parallel,  or  nearly  parallel,  to  the  axis  of  the 
cathode,  and  a  d-c  potential  applied  between  the  cathode  and  anode  sets  up  a  radial, 
static  electric  field.  Under  conditions  of  oscillation,  the  electrons  also  interact  with  an 
a-c  field. 

7.  THE  NON-OSCILLATING  MAGNETRON 

The  simplest  example  is  the  non-oscillating  solid  anode  magnetron.  Under  the  influence 
of  the  electric  field,  the  electron  is  impelled  to  move  toward  the  anode.  The  magnetic 
field  results  in  a  force  on  the  electron  which  is  normal  both  to  the  direction  of  motion  and 
to  the  direction  of  the  magnetic  field.  Theoretically,  there  is  a  minimum  critical  voltage, 
yc,  called  the  cutoff  voltage,  for  each  value  of  the  magnetic  field,  B,  for  which  electrons 
will  just  reach  the  anode.  The  formula  *  for  the  cutoff  voltage  is 


where  e  is  the  electronic  charge,  m  is  the  electronic  mass,  ra  is  the  anode  radius,  and  <r  is 
the  ratio  of  the  cathode  radius  to  the  anode  radius.  In  this  formula  it  is  assumed  that  the 
electrons  have  zero  velocity  at  the  cathode;  in  addition  the  relativistic  effects,  which 
become  significant  at  high  voltages,  are  neglected.  Since  Vc  is  proportional  to  Bz,  the 
locus  of  eq.  (1)  for  any  given  geometry  is  often  referred  to  as  the  cutoff  parabola. 

8.  THE  OSCILLATING  MAGNETRON 

Principal  current  interest  Hes  in  the  use  of  a  magnetron  as  a  self-excited  oscillator.  A 
convenient  classification  of  oscillating  magnetrons  can  be  made  by  distinguishing  among 
the  several  ways  in  which  electrons  interact  with  the  a-c  fields  to  sustain  oscillations. 
For  this  purpose,  three  types  of  interactions  can  be  identified. 

Type  I.  The  Negative  Resistance  Magnetron.  If  the  anode  of  the  magnetron  is  split 
into  halves  and  one  half  is  raised  to  a  higher  potential  than  the  other,  under  certain  condi- 
tions, most  of  the  electrons  will^go  to  the  plate  of  lower  potential.  In  this  event,  a  negative 

*  Unless  otherwise  indicated,  mks  units  will  be  used  in  articles  7-9. 


THE   OSCILLATING  MAGNETRON 


4-41 


resistance  exists  between  the  two  halves  of  the  anode,  and  oscillations  will  be  sustained  in 
a  tank  circuit  which  is  connected  between  them. 

Type  II.  The  Cyclotron  Frequency  Magnetron.  In  the  neighborhood  of  cutoff  (see 
eq.  [1]),  the  solid  anode  magnetron  will  sustain  oscillations  if  the  terminals  of  an  L-C  tank 
circuit  are  connected  between  the  cathode  and  anode.  The  wavelength  of  the  oscillations 
which  will  be  sustained  is  given  by 

\B  =  Constant  (2) 

The  most  commonly  observed  value  of  the  constant  is  about  15,000  centimeter-gausses. 
The  disadvantage  of  this  type  of  magnetron  is  that  the  electronic  efficiency  is  very  low. 


ROTATING  ANODE 
POTENTIAL   WAVE 


FIG.  1.     Approximate  Configuration  of  a  Space-charge  Cloud  of  an  8-segment  Magnetron  Operating 

in  the  7r-mode 

The  electrons  that  initially  absorb  energy  from  the  field  are  removed  from  the  interaction 
space  within  approximately  one  cycle  either  by  striking  the  anode  or  by  being  returned 
to  the  cathode.  Those  electrons  that  initially  yield  energy  to  the  field  stay  in  the  interac- 
tion space  for  a  longer  time.  However,  after  they  have  lost  their  energy  they  begin  to 
reabsorb  it  again  unless  some  method  is  provided  for  their  removal.  This  is  sometimes 
done  either  by  tilting  the  magnetic  field  slightly  with  respect  to  the  axis  of  the  cathode  or 
by  installing  electrodes  at  the  ends  of  the  interaction  space  which  are  made  positive  with 
respect  to  the  cathode.  In  either  method  the  electrons  that  have  given  energy  to  the  r-f 
field  are  drawn  off  at  the  ends  if  the  interaction  space  is  short  enough. 

Type  III.  The  Traveling  Wave  Magnetron.  In  this  magnetron,  the  anode  consists  of 
a  number  of  segments.  In  operation,  an  r-f  standing  wave  pattern  exists  in  the  interaction 
space  between  the  cathode  and  anode.  In  general,  standing  wave  patterns  may  be  thought 


4-42 


ELECTRON  TUBES 


FIG.  2o.    Schematic  Representation  of  a  Hole  and 
Slot  Magnetron 


of  as  being  composed  of  two  waves  traveling  in  opposite  directions.  The  standing  wave 
pattern  in  the  interaction  space  is  composed  of  two  traveling  waves  rotating  in  opposite 
directions  around  the  interaction  space. 

Oscillations  are  sustained  by  interaction  between  one  of  the  traveling-wave  components 
and  the  electronic  stream.  The  electronic  stream  assumes  the  shape  of  a  spoked  cloud 
which  is  centered  on  the  axis  of  the  tube  as  is  indicated  schematically  in  Fig.  1.  The 
spokes  wheel  around  the  interaction  space  in  synchronism  with  one  of  the  components  of 
the  rotating  wave  in  such  a  phase  that  the  spokes  are  in  a  retarding  tangential  *  electric 
field.  That  is  to  say  that  the  electrons  in  the  spokes  are  yielding  energy  to  the  r-f  field. 
Those  electrons  that  come  from  the  cathode  in  such  a  phase  as  initially  to  absorb  energy 

from  the  field  are  returned  to  the  cathode 
after  only  one  orbital  loop.  Figure  1  shows 
the  computed  paths  of  electrons  emitted 
at  several  different  phases.  The  paths 
are  drawn  as  they  would  appear  to  an 
observer  stationed  in  a  system  of  coordi- 
nates which  is  centered  at  the  axis  of  the 
tube  and  which  is  rotating  with  the  same 
angular  velocity  as  the  rotating  wave. 

This  selective  mechanism,  that  is,  the 
mechanism  by  which  the  unfavorable  elec- 
trons are  rejected  by  being  returned  to 
the  cathode  in  a  relatively  short  time  and 
the  favorable  electrons  are  grouped  into 
spokes  which  stay  in  a  retarding  r-f  field, 
results  hi  very  high  electronic  efficiencies. 
Electronic  efficiencies  of  60  per  cent  are 
not  uncommon. 

Associated  with  the  segments  of  the 
anode  is  a  system  of  resonators.  The 
resonator  system  may  assume  any  one  of  a 
variety  of  forms.  One  of  the  commonest 
forms  is  illustrated  in  Fig.  2a.  This  figure 
is  a  schematic  representation  of  the  hole- 
and-slot-type  magnetron.  Each  hole  and 
slot  can  be  thought  of  as  an  L-C  tank  cir- 
cuit with  an  associated  resonant  frequency. 
In  the  rising  sun  structure,  illustrated 
in  Fig.  26,  resonators  of  one  resonant  fre- 
quency alternate  with  resonators  of  an- 

other  frequency.     In   normal   operation, 

FIG.  25.    Schematic  Representation  of  a  Rising  Sun    *he  operating  frequency  of  the  ensemble 
Anode  Block  is  roughly  midway  between  the  resonant 

frequencies  of  the  two  sets. 

From  the  standpoint  of  practical  application,  the  traveling-wave  magnetron  is  by  far 
the  most  important  type  of  magnetron  oscillator.  It  will  be  the  exclusive  concern  of  article  9, 
and,  unless  explicitly  stated,  the  term  '  'magnetron"  will  mean  a  traveling-wave  magnetron. 


9.  OPERATION  OF  THE  TRAVELING-WAVE  MAGNETRON 

THE  R-F  PATTERNS  OF  THE  MODES.  In  a  magnetron  of  TV  segments  there  are 
N  possible  modes  of  oscillation.  The  different  modes  have  periodicities  of  n,  where  n  is 
any  of  the  integers  0,  1 ,  2,  •  •  •  N/2  if  N  is  even,  and  0,  1,2,  •  •  • ,  (N  -  1) /2  if  N  is  odd.  A 
mode  number,  or  designation,  is  numerically  equal  to  n.  Periodicity  is  here  defined  to 
mean  the  integral  number  of  repeats  of  a  field  pattern  in  a  single  revolution  around  the 
anode  at  any  given  instant. 

The  tangential  component  of  the  r-f  electric  field  is  zero  across  the  face  of  a  segment. 
Therefore,  the  field  distribution  of  a  given  mode  will  be  the  sum  of  an  infinite  series  of 
harmonics.  Each  harmonic  will  have  a  periodicity  of  k.  The  only  values  of  k  possible  are 
those  for  which 

k  =  n  -  pN  (3) 

where  p  is  any  integer — positive,  negative,  or  zero.    If  the  tube  is  of  the  rising-sun  variety, 
*  The  type  III  magnetron  is  also  called  the  tangential  resonance  magnetron. 


OPERATION  OF  THE  TRAVELING-WAVE  MAGNETRON      4-43 


m? 


?m    ??<??    ** 

II  ic—     '       ' 

mf 


w  w 


W 


FIG.  3.    Field  and  Potential  Distributions  of  the  Modes  in  an  8-segment  Magnetron 


4-44 


ELECTRON  TUBES 


where  the  resonators  which  connect  the  segments  are  alternately  large  and  small,  the 
allowed  values  of  k  will  be  given  by 

N 


n-p- 


(3a) 


That  is,  there  will  be  a  set  of  harmonics  associated  with  each  of  the  two  sets  of  resonators. 
The  mode  for  which  n  =  N/2  is  sometimes  called  the  TT  mode  for  the  reason  that  adja- 
cent segments  are  180°,  or  TT  radians,  out  of  phase. 

All  the  modes  except  the  zero  mode  and  the  TT  mode  occur  in  pairs,  or  doublets.  The 
two  members  of  a  doublet  have  the  same  periodicity  although  they  usually  differ  in  fre- 
quency and  the  patterns  are  displaced  with  respect  to  one  another  in  such  a  way  that  a 

current  loop   in    the    funda- 

24 '  ' '  '  '  '  '      mental  of  one  pattern  occurs 

in  the  same  position  as  a  cur- 
rent node  in  the  fundamental 
of  the  pattern  of  the  other. 

Some  of  these  principles  are 
illustrated  in  Fig.  3  for  an 
eight-segment  tube.  For  clar- 
ity only  the  electric  flux  lines 
of  the  fundamental  compon- 
ent are  shown  in  the  inter- 
action space.  To  illustrate 
the  field  configurations  in  the 
resonators,  only  the  magnetic 
flux  lines  are  shown.  Below 
these  are  plotted  the  distri- 
butions in  potential  for  the 
fundamental  component.  The 
reason  for  the  existence  of 
only  one  7r-mode  is  illustrated 
in  Fig.  3.  The  cos  40  solution 
corresponds  to  zero  potential 
on  all  the  segments. 

Although  the  frequencies  of 
the  several  modes  are  usually 
different,  all  the  harmonics 
of  a  given  mode  are  at  the 
same  frequency.  Thus,  if 
/n(  =  Wn/2-7r)  is  the  operating 
frequency  of  the  nth  mode, 
then  the  angular  velocity  of 
the  pth  harmonic  of  the  rotat- 
ing wave  is  un/k,  where  k  is 
given  by  eq.  (3) .  This  point 
is  stressed  for  the  reason  that, 
in  steady-state  operation,  the 
electronic  stream  interacts 
with  only  one  harmonic  of 
one  mode  at  any  instant. 
INPUT  CHARACTERISTICS.  The  d-c  voltage  for  which  an  electron  will  just  reach 
the  anode  for  an  infinitesimal  r-f  voltage  on  the  anode  of  frequency  /  and  periodicity  k  is 


\ 


1500. 


V 


\ 


\ 


30% 


CONSTANT   MAGNETIC  FIELD- GAUSS 

CONSTANT  POWER  OUTPUT- KILOWATTS 

—  -  CONSTANT  OVERALL  EFFICIENCY 
TYPICAL  OPERATING  POINT 


FIG.  4. 


16  20  24  28 

DC  CURRENT, I, IN  AMPERES 

Magnetron  Input  V-I  Characteristic 


32 


given  by 


VT 


k 


27r/m 


(4) 


Equation  (4)  defines  a  threshold  voltage.  Empirically  it  is  approximately  equal  to  the 
operating  voltage  at  low  currents  although  changes  in  the  r-f  loading  will  cause  changes  in 
the  input  d-c  voltage. 

For  any  fixed  loading  the  input  voltage  varies  approximately  linearly  with  input  cur- 
rent for  any  fixed  value  of  magnetic  field,  as  is  indicated  in  Fig.  4.  Figure  4  shows  voltage 
plotted  as  a  function  of  current  for  various  values  of  magnetic  field.  In  addition  to  the 
constant  magnetic  field  lines,  contours  of  constant  efficiency  and  constant  power  output 
are  also  shown.  A  method  for  obtaining  the  type  of  data  required  for  such  a  V-I  char- 
acteristic is  described  in  Section  11,  Microwave  Measurements. 


OPERATION   OF  THE  TRAVELING-WAVE   MAGNETRON      4-45 


Although  the  performance  chart  in  Fig.  4  is  typical,  it  does  not  indicate  the  possible 
range  of  operating  characteristics  and  parameters  of  magnetrons.  Table  1  lists  operating 
data  for  five  magnetrons.  The  data  for  each  tube  are  for  some  one  point  on  its  V-I  char- 
acteristic. The  most  striking  tiling  about  the  data  is  the  large  range  of  variation  of  the 
several  parameters  without  any  apparent  correlation.  It  will  be  shown  in  the  sequel, 
however,  that  the  data  do  fit  into  a  logical  scheme. 

Table  1 


Example 

Wavelength, 
centimeters 

Voltage, 
volts 

Current, 
amperes 

Magnetic 
field, 
gausses 

R-f  Power 
Output, 

Watts 

I 
II 
III 
IV 
V 

1 
3 
5 

10 
50 

15,000 
30,000 
2,500 
50,000 
2,000 

15.0 
40.0 
0,1 
200.1 
1.0 

8,000 
8,000 
3,000 
2,500 
800 

75,000 
500,000 
100 
4,000,000 
1,000 

SCALING.  If,  in  the  examples  in  Table  1,  all  the  tubes  were  assumed  to  be  working 
in  an  equivalent  way  (for  example,  all  with  the  same  electronic  efficiency),  then  the  data 
shown,  together  with  the  geometrical  parameters,  would  correlate  in  accordance  with  the 
principles  of  scaling.  The  principles  of  scaling  state  that  in  the  variation  of  the  parameters, 
7,  /,  B,  h,  7*a,  and  \  where  h  is  the  anode  length,  equivalent  operation  may  be  obtained 
by  maintaining  the  following  three  parameters  invariant : 

VB  (5) 

In  addition,  it  is  assumed  that  the  ratio  of  cathode  radius  to  anode  radius  and  the  number 
of  segments  also  are  maintained  constant. 

It  would  be  instructive  to  consider  an  example  of  the  type  of  sealing  in  which  every 
linear  dimension  is  scaled  by  the  same  factor,  S.  Let  the  unprimed  quantities  represent 
the  tube  from  which  it  is  desired  to  scale,  and  let  the  primed  quantities  represent  the  tube 
to  be  derived;  then 

V  =  SX 

ra'  =  Sra 
h'  =  Sh 

From  (5),  since  (X'AO  -  (XA)  and  (X'AV)  =  (X/r«), 

V'  =  V        I'  =  I        E'  —  — 

Thus,  the  new  tube  will  work  at  the  same  voltage  and  current  but  the  new  magnetic  field 
will  be  l/S  times  the  original  magnetic  field.  This  type  of  scaling  is  most  useful  when  the 
factor  S  does  not  differ  greatly  from  unity.  Ordinarily  it  cannot  be  applied  if  S  is  very 
large  or  very  small.  For  example,  suppose  that  X  =  30  cm  and  X'  =  3  cm.  In  this  case, 
$  —  0.1,  and,  since  I  =  I' ',  the  surface  current  density  at  the  cathode  will  be  up  by  a 
factor  of  100  in  the  new  tube.  A  current  density  of  10  amp  per  sq  cm  in  the  30-cm  tube 
is  a  quite  moderate  figure.  A  current  density  of  1000  amp  per  sq  cm  cannot  be  attained 
with  present  techniques.  In  order  to  scale  over  such  wide  ranges,  it  is  usually  necessary  to 
extend  the  scaling  laws  to  include  N  and  cr.  When  these  parameters  are  included,  the 
quantities  that  must  be  kept  invariant  are 

/i          \ 

a3)  (5a) 

where  k  is  used  instead  of  N  to  emphasize  the  importance  of  the  field  periodicity.  In 
7r-mode  operation  k  will  be  JV/2. 

Scaling  in  accordance  with  the  invariants  in  (5)  and  at  the  same  time  maintaining  the 
same  N  and  <r  gives  fairly  accurate  results.  Scaling  in  accordance  with  the  invariants  in 
(5a)  will  give  good  results  in  predicting  currents,  voltages,  and  magnetic  fields.  However, 
it  is  usually  found  that  equivalence  does  not  hold  for  large  changes  in  N  or  in  a:  In 
particular  it  is  usually  found  that  electronic  efficiency  decreases  with  increasing  N  and 
increasing  cr. 

MODE  SEPARATION  AND  MODING.  The  frequencies  of  the  several  modes  are 
separated  by  an  amount  which  depends  upon  the  type  and  degree  of  coupling  among  the 
several  resonators.  Such  mode  separation  is  advantageous  for  the  reason  that  it  allows 


4-46 


ELECTRON  TUBES 


(CO 


(b) 


1  r 

• 

1 

,                    fS 

"                                                                               "            SJ                                           " 

"    1 

I 

i 

,cz> 

ts 

I 

1 

«            *            „           '•           \  1                               ., 

r 

*.  \ 

SEC.  A-A 

SHOWING  LOCATION 

OF  STRAPS  IN  TOP 

OF  ANODE 


Cd> 


FIG.  5.   Magnetron  Anode  Strapping  Methods,    (a)  Early  British,  (6)  single  ring,  (c)  echelon,  (d)  double 

ring. 


OPERATION   OF  THE  TRAVELING-WAVE  MAGNETRON      4-47 

the  desired  mode  of  operation  to  be  excited  independently  of  the  other  modes.  This 
appears  to  be  one  of  the  conditions  necessary  for  high  electronic  efficiency. 

From  eq.  (3)  it  might  also  seem  desirable  to  separate  the  values  of  f/k  in  order  that  no 
two  modes  have  the  same  threshold  voltages.  This  has  appeared  to  be  true  in  some  cases, 
but,  in  general,  when  taken  alone,  this  condition  is  an  unreliable  index  of  the  possibility 
of  moding.  The  problem  is  complicated  by  several  factors  including,  principally,  such 
things  as  relative  r-f  loading  of  the  modes,  the  noise  levels  at  the  start  of  oscillations,  the 
transient  behavior  of  the  modulator  and  power  supply,  and  the  instability  of  the  space 
charge  of  the  magnetron  at  high  current  levels.  A  complete  analysis  of  moding  is  beyond 
the  scope  of  this  article.  However,  a  few  general  remarks  can  be  made.  In  these  remarks, 
it  will  be  assumed  that  N  is  even  and  the  7r-mode  operation  is  desired. 

Two  distinct  types  of  moding  have  been  observed.  One,  called  the  mode  skip,  occurs 
principally  in  the  high- voltage  pulse  magnetrons.  The  magnetron  fires  in  both  the  TT  mode 
and  an  unwanted  mode.  ("Unwanted  mode"  is  to  be  considered  as  being  denned  here  so 
as  to  include  a  non-oscillating  state  which  sometimes  occurs.)  However,  it  fires  in  only 
one  mode  during  any  one  pulse.  It  alternates  between  the  two  modes  in  more  or  less 
random  fashion.  This  type  of  moding  can  usually  be  cured  either  by  reducing  the  r-f 
loading  on  the  magnetron,  by  reducing  the  rate  of  rise  of  the  applied  voltage  pulse  or  by 
reducing  the  applied  voltage,  or  by  a  combination  of  the  three. 

A  mode  shift,  the  other  type  of  moding,  is  encountered  chiefly  in  low-voltage  c-w  mag- 
netrons. It  consists  of  a  shift  from  one  mode  to  another.  One  of  its  principal  causes  is 
an  inherent  instability  in  the  space  charge  at  high  current.  In  general,  it  can  be  cured  by 
reducing  the  r-f  loading  or  by  reducing  the  operating  current.  In  the  pulsed  case,  a  mode 
shift  is  usually  observed  to  occur  during  a  single  pulse  and  is  relatively  unaffected  by 
changes  in  the  rate  of  rise  of  the  applied  pulse.  If  the  cathode  is  well  designed,  the  high 
current  instability  will  occur  at  currents  which  are  lower  than  those  necessary  for  tempera- 
ture limitation.  Occasionally,  however,  mode  shifts  have  been  observed  which  involve 
instability  due  to  temperature-limited  operation. 

METHODS  OF  MODE  SEPARATION.  The  two  most  important  devices  for  achiev- 
ing frequency  separation  of  the  modes  involve  the  use  of  straps  and  of  the  rising-sun 
structure. 

Figure  5  shows  several  strapping  methods  schematically.  One  of  the  most  widely  used 
is  the  double  ring  strapping  of  Fig.  5a,  in  which  there  is  a  pair  of  concentric  rings  at  each 
end  of  the  anode.  One  ring  of  a  pair  is  connected  to  one  set  of  alternate  segments.  The 
other  ring  is  connected  to  the  other  alternate  set  of  segments.  In  -rr-mode  operation,  one 
strap  or  ring  is  everywhere  at  the  same  potential  and  the  two  rings  at  one  end  are  180° 
out  of  phase.  Hence,  the  straps  constitute  mainly  a  capacitance  loading  of  the  resonators 
with  a  resultant  increase  in  wavelength  in  the  TT  mode  over  the  unstrapped  case.  In  the 
n  =  1  mode,  the  potential  distribution  on  a  strap  is  almost  sinusoidal  and  periodic  in  only 
one  revolution  around  the  anode.  Moreover,  the  two  rings  in  a  set  are  at  nearly  the  same 
potential.  Thus,  the  principal  effect  of  straps  in  the  n  —  I  mode  is  a  shunt  inductance  so 
that  the  n  =  1  mode  wavelength  will  be  less  in  the  strapped  case  than  it  is  in  the  unstrapped 
case.  Modes  in  between  the  n  =  1  and  the  TT  mode  will  be  affected  in  a  way  intermediate 
between  the  two  extremes  first  mentioned.  That  is,  as  n  increases  from  1  to  TV/2,  the 
inductance  effect  of  the  straps  decrease  while  the  capacitive  effect  increases. 

Figure  6  shows  the  mode  spectra  of  three  different  types  of  18  segment  magnetrons. 
The  spectrum  in  Fig.  6a  is  for  an  unstrapped  symmetrical  tube,  that  in  Fig.  Qb  for 
a  strapped  symmetrical  tube,  while  Fig.  6c  shows  the  spectrum  for  a  rising-sun  tube. 

At  wavelengths  shorter  than  3  cm,  the  mechanical  problem  of  making  strapped  mag- 
netrons becomes  very  difficult.  The  rising-sun  structure  avoids  this  difficulty  by  provid- 
ing mode  separation  without  straps.  It  works  on  the  principle  that  the  normal  mode 
frequencies  of  a  system  of  resonators  become  separated  by  the  introduction  of  asymmetries. 
The  mode  spectrum  of  the  rising  sun  is  composed  of  two  branches.  Each  branch  is  asso- 
ciated with  one  of  the  two  sets  of  resonators. 

OUTPUT  COUPLING.  The  r-f  energy  can  be  coupled  to  the  load  in  several  ways.  Of 
these,  two  are  of  major  interest:  loop  coupling  and  wave-guide  coupling.  The  first  type  is 
indicated  schematically  in  Fig.  2a;  the  second  is  shown  schematically  in  Fig.  7.  In  Fig.  7, 
coupling  is  obtained  through  a  slot  in  the  back  of  one  of  the  resonators.  The  quarter-wave 
low-impedance  transformer  section  steps  down  the  impedance  of  the  wave  guide,  which  is 
of  the  order  of  all  ew  hundred  ohms,  to  the  low  impedance  required  at  the  slot  opening, 
which  is,  in  somejjiases,  as  low  as  1  or  2  ohms. 

The  choice  bew  teen  loop  output  or  wave-guide  output  is  usually  made  on  the  grounds 
of  mechanical  feMtbility  and  convenience.  At  wavelengths  greater  than  10  cm,  r-f  trans- 
mission is  ordinaJMy  done  in  coaxial  lines.  The  size  of  choke  assemblies  and  window  struc- 
tures in  wave  grilles  makes  them  cumbersome  and  difficult  to  fabricate.  For  these  rea- 


4-48 


ELECTRON  TUBES 


MODE  NUMBER, 


FIG.  6.    Mode  Spectra  for  Three  Different  Types  of  IS-segment  Magnetrons  Having  the  Same  7r-mode 
Wavelength,    (a)  Unstrapped  symmetrical  magnetron,  (Z>)  strapped  symmetrical  magnetron,  (c)  rising 

sun  magnetron. 


UNIFORM  OUTPUT  WAVEGUIDE  v 


TRANSFORMING 


FIG.  7.     Illustration  of  a  Method  of  Wave  Guide  Output  Coupling 


OPERATION   OF  THE  TRAVELING-WAVE  MAGNETRON      4-49 


sons,  loop  coupling  outputs  are  usually  found  at  the  longer  wavelengths.  Below  10  cm, 
wave  guides  are  usually  used  for  transmission,  and  the  physical  size  of  wave-guide  outputs 
is  reduced  to  manageable  proportions. 

In  common  with  any  self-excited  oscillator,  the  efficiency  and  frequency  stability  of  a 
magnetron  depends  upon  the  amount  of  coupling  between  the  magnetron  and  the  load.  In 
addition,  for  any  given  coupling,  the  efficiency  and  frequency  stability  will  change  as  the 
amount  of  loading  is  changed.  A  quantitative  measure  of  the  variation  of  efficiency  and 
frequency  with  loading  is  obtained  from  a  Rieke  diagram,  an  example  of  which  is  shown  in 
Fig.  8.  Contours  of  constant  power  output  and  contours  of  constant  frequency  are 


CONTOURS   OF  CONSTANT  POWER  OUTPUT 

CONTOURS   OF  CONSTANT   FREQUENCY 

FIG.  8.     Rieke  Diagram 

plotted  on  a  polar  diagram.  Load  impedances  on  a  polar  diagram  may  be  specified  by  a 
characteristic  impedance  of  the  transmission  line  into  which  the  magnetron  is  coupled, 
together  with  a  reflection  coefficient.  The  modulus  of  the  reflection  coefficient  is  propor- 
tional to  the  distance  from  the  pole  of  the  diagram  while  its  argument  is  proportional  to 
the  angular  displacement  around  the  diagram.  The  load  line  characteristic  impedance 
ZQ,  the  load  impedance  Z,  and  the  voltage  reflection  coefficient  r  are  related  by 


z  -  ZQ        v* 

f  -    rr     ,     ^     =   /^ 


(6) 


where  p  is  the  modulus  and  <j>  is  the  argument  of  r.  The  region  on  the  Rieke  diagram  of 
high  power  and  high  frequency  density  is  the  region  of  heavy  loading.  The  center  of  the 
diagram  corresponds  to  a  matched  load;  that  is,  the  load  impedance  is  equal  to  the  trans- 
mission-line characteristic  impedance.  Ordinarily,  the  loading  is  maintained  constant,  at 
the  match  point,  over  the  V-I  characteristic,  as  in  Fig.  4.  On  the  other  hand,  the  magnetic 
field  and  either  the  current  or  voltage  is  held  constant  over  a  Rieke  diagram.  A  descrip- 
tion of  some  of  the  techniques  for  obtaining  the  data  for  a  Rieke  diagram  and  a  discussion 
of  polar  diagrams  will  be  found  in  Section  11,  Microwave  Measurements. 

A  quantitative  measure  of  the  frequency  stability  with  respect  to  perturbations  in 
loading  at  the  match  point  is  the  prill  ing  figure.  Pulling  figure  is  defined  as  the  greatest 
excursion  of  frequency  observed  as  the  modulus  of  the  reflection  coefficient  is  maintained 
constant  at  a  value  of  0.2  and  its  phase  angle  is  varied  through  360°.  For  the  example 


4-50 


ELECTRON  TUBES 


shown,  the  pulling  figure  is  about  0.2  per  cent  of  the  center  frequency.  For  a  magnetron 
operating  at  a  wavelength  of  3  cm  this  would  represent  a  pulling  figure  of  20  Me. 

If  the  coupling  from  the  magnetron  to  the  transmission  line  is  made  tighter,  the  power 
output  and  the  pulling  figure  at  the  match  point  will  both  increase.  Nearly  all  magnetron 
output  circuits  are  designed  to  provide  some  predetermined  compromise  of  eflSciency  and 
pulling  figure  when  the  tube  is  operated  into  a  matched  line. 

TUNING.  The  frequency  of  oscillation  is  determined  almost  entirely  by  the  geometry 
of  the  cavities,  that  is,  by  the  equivalent  inductance  and  capacitance  of  the  oscillator  tank 
circuit.  In  order  to  tune  the  magnetron,  it  is  necessary  to  change  either  the  inductance 
or  capacitance  or  both. 

Figure  9  shows  three  tuning  methods  schematically.  The  inductive  pin  tuning  method 
is  depicted  in  (a) .  An  array  of  copper  pins  is  disposed  so  that  they  may  be  inserted  and 


r fr 


FIG.  9.    Methods  of  Magnetron  Tuning,    (a)  Inductive  pin,  (6)  segment-to-segment  capacitance,  (c) 

strap-to-strap  capacitance. 

withdrawn  from  the  inductive  portion  of  the  resonators.  As  the  pins  are  inserted,  the 
effective  inductance  is  decreased  and  the  frequency  goes  up.  Capacitance  variation 
schemes  are  shown  in  Figs.  9&  and  9c.  In  (6),  a  movable  conducting  ring  changes  capaci- 
tance between  the  segments.  In  (c),  the  ring  changes  the  capacitance  of  the  straps. 
The  schemes  shown  in  (a)  and  (c)  are  capable  of  giving  tuning  ranges  of  about  20 
per  cent.  The  scheme  shown  in  (6)  is  usually  limited  to  a  somewhat  shorter  tuning 
range  for  the  reason  that  different  modes  tune  at  such  widely  varying  rates  with 
respect  to  tuner  displacement  that  the  TT  mode  tunes  but  a  relatively  short  distance  before 
encountering  interference  from  other  modes. 


BIBLIOGRAPHY 

Collins,  G.  B.  (ed.),  Microwave  Magnetrons,  M.I.T.  Radiation  Laboratory  (sponsor).    McGraw-Hill 

(1947). 

Brainerd,  J.  G.f  et  aZ.,  Ultra-High  Frequency  Techniques.  Van  Nostrand  (1942). 
Fisk,  Hagstrum,  and  Hartman,  Bell  Sys.  Tech.  J.,  Vol.  25,  167  (1946). 
Herriger  and  Hulster,  Zeit.  /.  Hockfrequenz.,  Vol.  49,  123  (1937). 
Kilgore,  G.  R-,  Proc.  I.R.B.,  Vol.  24,  1140  (1936). 
Postnumus,  K.,  Wireless  Engineer,  VoL  12,  126  (1935). 
Megaw,  E.  C.  S.,  J.  I.E.E.  (London),  Vol.  72,326  (1933). 
Okabe,  K,  Proc.  I.R.E.,  VoL  17,  652  (1929). 
Hull,  A.  W.,  Phys.  Rev.,  Vol.  18,  31  (1921). 
Habann,  Zeit.f.  Hochfrequenz.,  Vol.  24,  115  and  135  (1924). 
Blewett  and  Ramo,  Phys.  Reo.t  Vol.  57,  635  (1940). 


KLYSTRONS   (EMPLOYING  TRANSIT  TIME  BUNCHING)      4-51 


KLYSTRONS 

By  A.  L.  Samuel 

Electron  tubes  which  make  use  of  the  principle  of  velocity  modulation  are  now  known 
as  klystrons.  The  name  klystron  was  originally  a  registered  trademark.  It  is  now  applied 
to  all  tubes  of  the  same  general  type  without  regard  to  the  manufacturer.  A  klystron 
has  been  denned  by  the  IRE  as  an  electron  tube  in  which  the  distinguishing  features  are 
the  modulation  or  periodic  variation  of  the  longitudinal  velocity  of  an  electron  stream 
without  appreciable  variation  of  its  convection  current  and  the  subsequent  conversion  of 
this  velocity  modulation  into  convection-current  modulation  by  the  process  of  punching. 
All  commercial  tubes  of  the  klystron  type  make  use  of  cavity  resonators,  although  such 
resonators  are  not,  in  principle,  essential  to  their  operation.  Two  types  of  klystrons  are 
in  general  use:  tubes  of  the  first  type  bear  no  further  designation:  tubes  of  the  second 
type  are  referred  to  as  reflex  klystrons  or  simply  reflex  tubes. 


10.  EXYSTRONS  (EMPLOYING  TRANSIT  TIME  BUNCHING) 

A  few  typical  klystrons  are  shown  in  Figs.  1  and  2.  As  already  stated,  it  is  customary, 
although  not  essential,  to  employ  resonant  cavities  as  the  tuned  circuits  associated  with 
the  input  and  output  portions  of  these  amplifiers.  These  cavi- 
ties take  the  place  of  conventional  circuits  and  must  be  tuned 
to  the  operating  frequency.  They  may  be  partly  external  to 
the  tube  proper,  or  they  may  form  an  integral  part  of  the  tube 
as  supplied  to  the  user.  Cavities  are  used  because  they  produce 
larger  effective  fields  in  the  interaction  gap  regions  than  could 
be  obtained  by  any  other  means. 

The  basic  principles  of  the  klystron  amplifier  may  be  ex- 
plained by  referring  to  Fig.  3.  This  figure  illustrates  a  tube 
which  consists  of  (1)  an  electron  gun,  composed  of  a  heater,  a 
cathode,  and  auxiliary  focusing  electrodes;  (2)  an  input  region 
called  the  input  gap,  defined  by  two  grids  which  in  this  case 
form  a  part  of  a  cavity  resonator;  (3)  a  conversion  region  called 
the  drift  space  which  is  relatively  free  of  electric  or  electro- 
magnetic fields;  (4)  an  output  region,  called  the  output  gap, 
again  defined  by  two  grids  which  form  a  part  of  a  second  cavity 
resonator;  and  (5)  a  collector  electrode  whose  sole  function  is  to 
collect  the  electron  stream  after  it  has  traversed  the  working 
region  of  the  tube.  These  five  portions  of  the  tube  correspond 
directly  to  the  five  essential  operations  that  must  be  performed 
in  any  electron  tube.  The  separation  of  these  operations  makes 
it  possible  to  consider  them  separately  and  to  explain  the  opera- 
tion of  the  device  in  very  simple  terms.  These  operations  are, 
obviously,  (1)  the  production  of  an  electron  stream,  (2)  the 
modulation  or  variation  of  some  property  of  this  stream  in  ac- 
cordance with  an  input  signal,  (3)  the  conversion  of  the  original  modulation  into  a  form  in 
which  it  can  be  utilized,  (4)  the  utilization  of  the  stream  to  produce  an  output  signal,  and 
(5)  the  collection  of  the  electron  stream. 

Referring  to  Fig.  3,  the  field  in  the  input  gap  region  of  the  tube  varies  the  velocity  of 
the  electron  stream  in  a  cyclic  manner,  the  variation  in  velocity  being  assumed  to  be  small 
compared  to  the  average  velocity  imparted  to  the  stream  by  the  d-c  fields.  Those  electrons 
which  arrive  when  the  field  is  in  an  aiding  direction  are  speeded  up;  those  arriving  a  half 
cycle  later  are  slowed  down.  The  contributions  in  energy  made  by  the  field  to  some  of  the 
electrons  of  the  stream  is  nearly  balanced  by  the  energy  taken  from  those  electrons  which 
are  slowed  down.  The  modulating  process,  therefore,  requires  substantially  no  power, 
most  of  the  input  power  being  consumed  by  ohmic  losses  in  the  walls  of  the  input  cavity. 
All  the  electrons,  except  those  intercepted  by  the  grids,  proceed  through  the  next  region 
of  the  tube,  the  so-called  drift  space,  where  the  electron  stream  becomes  bunched  through 
the  simple  process  of  the  faster  electrons,  that  is,  those  that  are  speeded  up  by  the  field  in 
the  input  gap,  overtaking  the  group  of  slower  electrons  that  precede  them.  This  bunching 
process  converts  the  original  velocity  modulation  into  a  variation  in  the  rate  at  which 
electrons  pass  any  given  point.  The  stream  as  it  crosses  the  output  gap  appears  super- 
ficially like  the  stream  of  electrons  in  the  screen-grid  plate  region  of  the  conventional  space- 


la,    A  Typical  Klys- 
tuon  of  the  Integral  Cavity 
Type    Shown    without    Its 
Tuner.      (The   3K30   oscil- 
lator amplifier.) 


4-52 


ELECTRON  TUBES 


Coarse  Tuning 
Adjustment  • 


Tuning  Ring 
Clamp 


Loading  Spring 


Coarse 
Tuning 
Adjustment 


Tuner  Knob 


Loading 
Spring 
Spreading 

Cone        Tuner  Struts 


FIG.  16.     The  3K30  Shown  in  Section  with  an  11-C  Tuner  Attached 

charge  control  tetrode.  It,  therefore,  induces  currents  in  the  output  cavity  and  delivers 
power  to  the  output  in  just  the  same  way  that  an  electron  stream  delivers  energy  to  the 
field  between  the  screen  and  the  plate  of  the  conventional 
tube.  The  tuning  of  the  output  cavity  must  be  adjusted  to 
cause  the  maximum  value  of  the  field  to  occur  in  a  retarding 
direction  at  the  time  that  the  effective  center  of  an  electron 
bunch  crosses  the  output  gap.  Finally,  the  spent  electron 
stream  is  collected  by  a  final  electrode  where  the  energy  re- 
maining in  the  stream  is  dissipated  as  heat.  This  is  to  be 
contrasted  with  the  ac- 
tion in  the  conventional 
tube  where  the  plate 
performs  the  dual  func- 
tion of  providing  the 
output  circuit  retarding 
field  and  of  dissipating 
unused  energy  as  heat. 

An  interesting  possi- 
bility exists  in  the  klys- 
tron amplifier  of  utilizing 
the  electron  stream  in 
cascade  to  provide  either 
a  multistage  amplifieror 
a  combination  function 
device  such,  for  example, 
as  an  oscillator  buffer 


—        -^r,— -  Collector 


•Output  Gap 


Input 
Terminal 

Input 
Resonator 


•Drift  Space 
Input  Gap 


'Electron  Gun 


FIG.  2.   The  Type  2K47  Klys- 
tron. Frequency  Multiplier 


FIG.  3.    A  Sectional  View  of  a  Klystron 
Amplifier 


amplifier.  When  the  electron  stream  traverses  the  output  gap  in  the  simple  amplifier,  it 
obtains  an  augmented  velocity  modulation  as  a  result  of  the  higher  field  intensity  existing 
in  this  cavity — this  at  the  same  time  that  the  stream  delivers  energy  to  the  cavity  because 
of  its  bunched  condition.  This  augmented  modulation  is  in  quadrature  with  the  original 


KLYSTRONS    (EMPLOYING  TRANSIT  TIME  BUNCHING)       4-53 

modulation.  At  low  modulation  levels,  such  as  those  obtained  in  an  amplifier,  it  can  be 
thought  of  as  existing  quite  independently  of  the  original  modulation.  In  the  ordinary 
two-cavity  single-stage  amplifier,  no  use  is  made  of  this  added  modulation.  However,  by 
providing  a  second  drift  space  and  a  third  cavity,  an  additional  stage  of  amplification  can 
be  obtained.  The  middle  cavity  or  cascade  cavity  need  have  no  external  connection,  al- 
though, if  broad  band  amplification  is  desired,  it  is 
necessary  to  load  this  cavity  in  some  fashion  to  reduce 
its  effective  Q  to  a  value  comparable  to  that  of  the 
input  and  output  cavities  which  are  loaded  by  their 
external  circuit  connections. 

The  gain  of  a  klystron  amplifier  varies  with  the 
input  signal  level  in  quite  a  different  way  from  the 
behavior  of  other  types  of  amplifiers.  At  low  levels, 
the  gain  depends  only  on  the  beam  current,  the  beam 
voltage,  and  the  physical  dimensions  of  the  tube. 
However,  as  the  drive  is  increased  beyond  a  certain  Input 

point,  the  phenomenon  of  overbunching  sets  in  and     FIG.  4.  The  Variation  in  Output  Power 
the  gain  begins  to  decrease.    Eventually  a  maximum     *lth  tne      o4rbu^hmg 
output  is  reached ;  with  a  further  increase  in  the  input, 

the  output  actually  decreases.  This  is  illustrated  in  Fig.  4,  where  the  output  as  a  function 
of  the  input  is  plotted  for  a  typical  klystron  amplifier. 

At  low  levels,  the  gain  of  a  klystron  amplifier  is  given  by 

Gain  «  ZiZ2Mi*M22Sz  (1) 

where  Zi  and  Z%  are  the  impedances  of  the  input  and  output  cavities  respectively  as  measured  across 
the  interaction  gaps.  The  parameters  Mi  and  M-i  are  the  absolute  values  of  the  input  and  output 
beam  coupling  coefficients  or  so-called  modulation  coefficients  and  express  the  effectiveness  with  which 
the  fields  in  the  cavities  interact  with  the  electron  stream.  The  parameter  5  is  the  absolute  value  of 
the  beam  transadmittance  and  is  a  measure  of  the  ratio  of  current  variation  produced  by  the  bunching 
process  to  the  equivalent  voltage  variation  impressed  on  the  beam. 

The  cavity  impedances  can  be  measured  directly  if  desired,  or  they  may  be  computed  from  measured 
values  of  the  characteristic  impedance  and  Q  of  the  cavity. 

The  absolute  value  of  the  beam  coupling  coefficient  for  the  gap  between  two  axial  cylinders 
without  grids  varies  for  different  electrons  depending  upon  their  distance  from  the  axis  of  the  beam. 
The  value  for  that  portion  of  the  beam  lying  at  a  radial  distance  from  the  axis  of  8p  where  6p  is  expressed 
in  radians  at  the  operating  frequency  and  with  tubes  having  a  radius  of  Br  (also  in  radians)  is  given  by 

^  =  !inJ^2).^M  (2) 


where  the  Jo's  are  modified  Bessel  functions  of  the  first  kind. 

The  absolute  value  of  the  beam  coupling  coefficient  for  an  interaction  gap  between  ideal  grids  is 
given  by 


where  9g  is  the  electron  transit  angle  of  the  gap  denned  as  the  time  required  for  the  unmodulated  electron 
beam  to  cross  the  gap  measured  in  radians  at  the  operating  frequency.  The  value  of  6  may  be  com- 
piled from 


, 

XV? 

where  X  is  the  gap  spacing,  X  is  the  free  space  wavelength  corresponding  to  the  operating  frequency, 
and  V  is  the  voltage  corresponding  to  the  velocity  of  the  electron  beam.  The  parameters  X  and  X 
are  measured  in  the  same  units  (usually  centimeters)  ,  and  V  is  in  volts. 

The  absolute  value  of  the  beam  transadmittance  for  low  signal  levels  is  given  by 

fl-£'  C5) 

where  8  is  the  electron  transit  angle  in  the  drift  space,  I  is  the  beam  current  in  amperes,  VQ  is  the 
beam  voltage,  and  <r  is  a  reduction  factor  to  account  for  certain  space-charge  effects  that  will  not  be 
discussed.  In  well-designed  amplifiers,  the  parameter  <r  is  usually  of  the  order  of  0.5  at  the  recommended 
operating  conditions  and  increases  to  1.0  as  the  beam  current  (at  a  given  voltage)  is  decreased  to  a 
low  value. 

At  high  levels  the  beam  trans  conductance  is  sometimes  expressed  as 

where  V  is  the  value  in  volts  of  the  velocity  modulation  impressed  on  the  beam,  and  */i  is  a  Bessel 
function  of  the  first  kind. 

This  expression  is  based  on  kinematic  considerations  only  and  neglects  space  charge  and  other 
sources  of  non-linearity.  It  may  be  used  as  a  rough  basis  for  predicting  the  general  behavior  of  a, 
klystron  amplifier  for  large  signals,  but  it  does  not  agree  quantitatively  with  experimental  results. 


4-54 


ELECTRON  TUBES 


11.  REFLEX  KLYSTRONS 

A  reflex  tube  or  reflex  klystron  is  a  special  form  of  the  klystron  oscillator  employing  a 
single  cavity  with,  a  single  interaction  gap  to  perform  the  functions  of  both  the  input  and 

output  circuits  (see  Figs.  5  and  6).  The 
electron  stream  is  velocity  modulated  on  a 
first  transit  of  this  gap  and  is  forced  to 
cross  the  gap  a  second  time  by  means  of  a 
repelling  or  reflecting  field.  The  electrons 
become  bunched  in  the  process  of  reflection, 
the  speeded-up  electrons  penetrating  the 
field  to  a  great  distance  and  therefore 
taking  longer  to  return  than  the  slowed- 
down  electrons  in  just  the  same  way  that 
a  ball  thrown  upward  in  the  earth's  gravi- 
tational field  takes  longer  to  return  if 
thrown  with  high  velocity  than  if  thrown 
with  low  velocity.  In  order  for  the  oscilla- 
tions to  be  self-sustained,  the  returning 
bunches  of  electrons  must  arrive  at  the 
interaction  gap  at  the  correct  phase  of  the 
alternating  field,  that  is,  when  the  field  has 
its  maximum  value  in  the  retarding  direc- 
tion. This  requires  that  the  electrons  re- 
main in  the  reflecting  field  region  for  a 
critically  valued  length  of  time,  a  time 


FIG.  5.    The  707A  Reflex  Klystron  Em- 
External  Cavity,  Shown  with  the  Ca- ' 
Disassembled 


nploying  t._ 

that  is  approximately  n  H-  3/4  cycles  at 
the  operating  frequency,  where  n  is  any 
integer  equal  to  or  greater  than  zero.  This  time  can  be  adjusted  by  varying  the  velocity 
of  the  beam  as  it  crosses  the  gap  on  the  first  transit  or,  more  usually,  by  varying  the 
voltage  of  the  repeller.  As  the  voltage  of  the  repeller  is  varied,  a  series  of  operating  regions 


Resonator. 
CoopJIpg  Loo| 
Flexible  Diaphragm 

Tuner  S'cre'w.- 

Accelerating  Grid- 
Tuner  Bow- 


Tuner  Back  Strut 
Repeller 

•Cavfty  Grfds 
Beam  Forming 
Electrode 
Cathode 
Cathode  Heater 


Coaxial  Output  Lead- 


FIG.  6.    The  2K25  Reflex  Klystron,  a  Mechanically  Tuned  Tube  of  the  Integral  Cavity  Type 


called  modes  will  be  observed  corresponding  to  different  values  of  n  in  the  above  relation- 
ship. Only  a  limited  number  of  modes  corresponding  to  values  of  n  from  1  or  2  to  4  or  5 
are  actually  observed,  and  usually  only  one  or  two  of  these  modes  produce  enough  power 
to  be  useful.  I'he  variation  in  the  output  power  for  a  typical  reflex  tube  is  shown  in  Fig.  7. 
It  will  be  observed  that  oscillations  are  actually  produced  for  a  limited  range  in  voltage 
in  the  vicinity  of  the  optimum  values,  and  that  a  variation  of  frequency  occurs  as  shown 
by  the  top  curves.  This  variation  in  frequency  with  voltage  is  called  electronic  tuning. 


REFLEX  KLYSTRONS 


4-55 


Electronic  tuning  is  often  employed  as  a  means  for  critically  adjusting  the  operating 
frequency  in  applications  where  the  accompanying  variations  in  the  output  power  can  be 
tolerated,  such,  for  example,  as  a  local 
oscillator  in  a  superheterodyne  receiver. 
Electronic  tuning  finds  its  greatest  useful- 
ness in  connection  with  automatic  fre- 
quency control  circuits.  Since  the  mode 
with  the  highest  output  has  the  smallest 
electronic  tuning  band  width,  a  compromise 
must  often  be  made  between  output  and 
tuning  range.  The  electronic  tuning  range 
is  generally  small  as  compared  to  the  usual 
mechanical  tuning  range.  When  electronic 
tuning  is  employed,  it  is  essential  that  the 
mechanical  tuning  be  adjusted  so  that  the 
operating  point  falls  somewhere  near  the 

middle  of  the  electronic  tuning  range  where     -700  -soo  —soo  —400  -300  -200  -100 
the  frequency  can  be  shifted  a  reasonable  Reflector  Voltage 

amount  in  either  direction  with  the  elec-  FIG.  7.  A  Typical  Mode  Curve  for  a  Reflex  Elys- 
tronic  control  without  too  much  cha^e  in 
output  power.  Electronic  tuning  and  other 
associated  phenomena  can  be  explained  by 
considering  the  way  in  which  the  impedance  of  the  electron  stream  as  seen  by  the  cavity 
varies  with  the  electron  transit  time  in  the  repeller  region. 

Figure  8  is  a  plot  of  the  small  signal  beam  conductance  showing  the  relationship 


1 

i 

-f-20  ° 

\ 

J.1Q    ag 

N 

\ 

\ 

o    '8- 

\ 

\ 

\ 

\ 

V 

\ 

si 

Be 

im 

vol 

12 

50 

i 

en 

<v 

DltS 

ge 

- 

40 

~30U* 

1 

f  \ 

Fr« 

qu< 

nc 

ft 

4c) 

92 

50 

80=~ 

1 

i 

/• 

00^ 

1 

\ 

/ 

60   0     = 

I 

\ 

/ 

A 

1 

1 

f 

/\ 

o  •* 

the  2K41  Tube) 


= 
* 


(VM6/2V0') 


2V  Q       VM6/Vo 


(7) 


Negative  of 
Circuit  Admittance 

Small  Signal  Admittance 
of  Electron  Stream. 


where  the  first  term  represents  the  magnitude  of  small  signal  admittance,  and  the  last  term  is  the 
phase  angle.    Here  0  is  the  transit  angle  in  the  repeller  region,  and  the  rest  of  the  symbols  have  the 

same  significance  as  in  the  article 
on  klystrons  employing  transit 
time  bunching.  The  second  term 
accounts  for  the  decrease  in  mag- 
nitude of  the  conductance  which 
occurs  for  large  signal  levels  and 
is  similar  to  the  compression  term 
appearing  in  the  expression  for 
the  small  signal  transadmittance 
of  the  klystron  amplifier.  The 
final  term  gives  the  phase  angle 
of  the  conductance. 

If  the  real  part  of  this  admit- 
tance is  negative  and  larger  in 
magnitude  than  the  positive  real 
component  of  the  cavity  admit- 
tance, oscillations  will  build  up 
until  limited  by  the  non-linearity 
given  by  the  second  term.  Under 
stable  operating  conditions,  the 
relationship 

Yg  +  Ye  _  0  (8) 

must  be  satisfied,  where  Yc  is  the 
admittance  of  the  cavity.  The 
cavity  may  be  assumed  to  behave 
as  a  simple  shunt  tuned  circuit 
in  the  vicinity  of  the  resonant 
frequency  so  that 

FIG.  8.     The  Admittance  of  the  Electron  Stream  as  Viewed  from  the  .    /  . 

Cavity  for  a  Reflex  Klystron.    Oscillations  can  occur  only  in  the  re-    yc  =  G  +  —  I  --  !  —  -  1       (9) 
gion  where  the  negative  conductance  of  the  beam  exceeds  the  pos-  QQ  VWQ        w  / 

itive  conductance  of  the  circuit. 


Typical  Point  at  which 
Oscillations  may  Start, 


ance,  QQ  is  the  cavity  Q,  «Q  is  the  angular  frequency  at  resonance,  and  a  is  the  angular  frequency  cor- 
responding to  the  particular  value  of  Yc.  The  negative  of  this  value  is  plotted  in  Fig.  8  and  appears 
as  the  straight  line  to  the  left  of  the  imaginary  axis. 

Oscillations  will  not  be  sustained  for  all  values  of  Yg  lying  in  the  shaded  area  on  the  figure.  For  values 
of  Yg  lying  to  the  left  of  the  Yc  line,  oscillations  will  build  up  until  Yg  sinks  along  a  radial  line  arriving 
at  a  stable  operating  point  on  this  line.  If  the  operating  point  lies  on  the  real  axis,  the  oscillations  vriH 
occur  at  the  resonant  frequency  of  the  cavity.  Operating  points  off  the  axis  correspond  to  oscula- 
tions at  a  frequency  which  differs  from  the  resonant  frequency  by  a  sufficient  amount  to*  provide 


4-56 


ELECTRON  TUBES 


the  necessary  reaction  component  of  admittance  specified  by  the  operating  point.    This  effect  is  called 
electronic  tuning. 

The  output  impedance  characteristic  of  a  reflex  tube  can  best  be  illustrated  by  plotting 
this  characteristic  on  the  reflection  qoefficient  plane  (sometimes  called  a  Rieke  diagram  on 
a  Smith  chart) .  A  typical  plot  is  shown  in  Fig.  9,  where  lines  for  constant  power  are  shown 
solid  and  lines  for  constant  frequency  are  plotted  on  a  background  of  orthogonal  circles 
representing  fixed  values  of  the  resistive  and  reactive  components  of  the  load  impedance. 
The  region  on  the  plot  where  the  constant-frequency  lines  tend  to  converge  is  called  the 


ERes<=  300  Volts 

FIG.  9.    The  Variation  in  Output  Power  and  Frequency  with  the  Load  Impedance  for  a  Typical  Reflex 
Klystron  (the  2K25)  Shown  on  the  Reflection  Coefficient  Plane 

frequency  "sink,"  and  the  minimum  amplitude  of  standing  wave  ratio  that  will  cause  the 
tube  to  operate  in  this  region  of  discontinuity  is  called  the  "sink  margin."  It  is  customary 
to  require  a  sink  margin  of  8  db.  A  second  important  characteristic  is  the  so-called  pulling 
figure  which  is  defined  as  the  maximum  difference  in  frequency  produced  when  a  mismatch, 
having  a  reflection  coefficient  of  0.2  as  measured  at  the  prescribed  output  coupler,  is 
varied  through  360°.  The  pulling  figure  for  the  tube  shown  in  Fig.  9  is  approximately  4  Me. 

The  power  output,  electronic  tuning  sink  margin,  and  pulling  figure  for  the  typical  reflex 
tube  will  be  found  to  vary  somewhat  over  the  mechanical  or  thermal  tuning  range  of  the 
tube.  Curves  showing  these  variations  for  any  particular  tube  are  customarily  supplied 
in  the  technical  information  sheets  published  by  the  manufacturers, 

The  coarse  adjustment  of  frequency  is  usually  made  by  mechanical  means,  either  by 
varying  the  effective  size  of  the  external  portion  of  the  cavity  or  by  internal  changes, 
usually  of  the  length  of  the  interaction  gap,  and  hence  of  the  effective  capacitance  loading 
of  the  cavity.  In  most  tubes  where  capacitance  tuning  is  employed,  the  necessary  motion 
is  transmitted  through  the  vacuum  envelope  by  means  of  a  flexible  diaphragm.  Recently, 
a  number  of  tubes  have  been  introduced  in  which  mechanical  cavity  tuning  is  produced 


TUBE  TYPES 


4-57 


internally  by  thermal  expansion  means.  This  method,  called  thermal  tuning,  makes  it 
possible  to  adjust  the  frequency  of  the  tube  over  its  entire  mechanical  tuning  range  by 
electrical  means,  without  the  large  variations  in  output  power  that  are  encountered  with 
electronic  tuning.  The  thermal  tuning  speed  is  limited  by  the  thermal  capacity  of  the 
tuning  mechanism,  but  it  is  sufficiently  fast  for  many  automatic  frequency  control  appli- 
cations. 

12.  TUBE  TYPES 

Many  of  the  tubes  listed  in  Table  1  were  made  for  the  armed  services  during  World 
War  II,  and  some  may  not  be  commercially  available.  The  prospective  user  should  con- 
sult the  manufacturer  in  regard  to  their  availability  and  should  follow  his  recommenda- 
tions regarding  operating  conditions  and  ratings.  The  data  of  Table  1  are  indicative  of 
typical  operating  conditions  and  are  supplied  for  general  reference  purposes  only. 

Table  1.    Western  Electric  Reflex  Tubes — External  Cavity  Type 


Number 

Frequency, 
megacycles 

Reso- 
nator 
Volt- 
age 

Reflector 
Voltage 
(Negative) 

Heater 
Volt- 
age 

Out- 
put, 
milli- 
watts 

Tuning  Range, 
megacycles 

Remarks 

Mechani- 
cal 

Elec- 
tronic 

Ther- 
mal 

707A/B 
2K48 

2,500-  3,750 
3,000-10,000 

300 
1,000 

0-    275 
0-    500 

6.3 
6.3 

70 
24 

* 

35 
10 

Table  1 — Continued.    Western  Electric  Reflex  Tubes — Internal  Cavity  Type 


7260 
726B 
726A 
2K29 
2K56 
2K54 
2K23 
2K55 
2K22 
2K26 
2K25 
2K45 
723A/B 
2K50 

2,700-  2,930 
2,880-  3,170 
3,170-  3,410 
3,400-  3,960 
3,840-  4,460 
4,290-  4,560 
4,275-  4,875 
4,590-  4,860 
4,240-  4,910 
6,250-  7,060 
8,500-  9,660 
8,500-  9,660 
8,700-  9,550 
23,215-24,750 

300 
300 
300 
300 
300 
1,130 
1,130 
1,130 
300 
300 
300 
300 
300 
300 

50-    210 
50-    210 
50-    210 
50-    210 
125-    175 
800-1,100 
600-    900 
625-    850 
75-    235 
70-    150 
75-    200 
95-    145 
90-    200 
20-    130 

6.3 
6.3 
6.3 
6.3 
6.3 
6.3 
6.3 
6.3 
6.3 
6.3 
6.3 
6.3 
6.3 
6.3 

120 
120 
120 
95 
65 
700  t 
250  f 
700  t 
85 
50 
30 
30 
25 
10 

230 
290 
240 
560 
620 
270 
600 
270 
670 
810 
1,160 

25 
25 
25 
32 
30 

Pulsed 
Pulsed 
Puked 

8  sec  tuning  time 
2  sec  tuning  time 

35 
35 
32 
50 
32 
65 

1,160 
935 

850 

The  data  above  were  obtained  on  tubes  manufactured  for  the  Army  and  Navy. 

Table  1 — Continued.    Raytheon  Manufacturing  Company 


T?aon 

Out- 

Tuning Range, 

Type 

Frequency, 
megacycles 

iteso- 
nator 
Volt- 

Reflector 
Voltage 
(Negative) 

Heater 
Volt- 
age 

put 
Power, 
milli- 

megacycles 

Remarks 

Mechan- 

Elec- 

Ther- 

age 

watts 

ical 

tronic 

mal 

QK269 

1,200-  1,500 

300 

100-220 

6.3 

150 

300 

12 

707B 

3,400-  3,600* 

300 

155-290 

6.3 

150 

20 

Requires  external 

cavity 

2K28 

3,400-  3,600* 

300 

155-290 

6.3 

150 

20 

Requires  external 

cavity 

QK159 

2,950-  3,250 

300 

112-250 

6.3 

150 

20 

5721 

4,290-  8,340* 

1000 

60-600 

6.3 

160 

12 

Requires  external 

(min.) 

cavity 

2K25/723A-B 

8,500-  9,660 

300 

85-200 

6.3 

33 

1160 

40 

2K33 

23,710-24,290 

1800 

80-220 

6.3 

40 

580 

40 

Table  1 — Continued.    Sylvania  Electric  Products,  Inc. 


6BL6 

1,600-  5,500* 

350 

15-700 

6.3 

125 

10 

Requires  external 

cavity 

6BM6 

500-3,000* 

350 

15-700 

6.3 

60 

13.4 

Requires  external 

cavity 

*  Tuning  range  of  external  cavity  tubes  depends  upon  cavity  design  and  may  be  anything  up  to  the  total  range  of 
the  tube. 

f  Average  power  based  on  0.1  duty. 


4-58 


ELECTRON  TUBES 


Table  1 — Continued.    Sperry  Gyroscope  Company — Reflex  Klystrons 


Type 

Frequency, 

Resonator 

Vn1+«aer« 

Reflector 
Voltage 

Heater 

Grid 

Vnl+au'p 

Output 

Tuning  Range, 
megacycles 

(Negative) 

Mechanical 

Electronic 

3K27 

750-      960 

,000 

0-1,500 

6.3 

+20  to  -200 

I  w 

210 

10 

3K23 

950-  1,150 

,000 

0-1,500 

6.3 

+20  to  -200 

1  w 

200 

10 

2K41 

2,660-  3,310 

,250 

0-    750 

6.3 

+50  to  -200 

250  mw 

650 

17 

2K42 

3,300-  4,200 

,250 

0-    750 

6.3 

+30  to  -200 

250  mw 

900 

15 

2K43 

4,200-  5,700 

,250 

0-    750 

6.3 

+30  to  -200 

250  mw 

1,500 

15 

2K44 

5,700-  7,500 

,250 

0-    750 

6.3 

+30  to  -200 

250  mw 

1,800 

15 

2K39 

7,500-10,300 

,250 

0-    750 

6.3 

+30  to  -200 

250  mw 

2,800 

44 

Table  1 — Continued.    Sperry  Gyroscope  Company — 2-cavity  Oscillator/ Amplifiers 


Type 

Frequency, 
megacycles 

Reso- 
nator 
Volt- 
age 

Grid 
Voltage 

Heater 
Volt- 
age 

Output 

Tuning  Range,  megacycles 

Remarks 

Mechanical 

Elec- 
tronic 

3K2I 
3K30/ 
410R 
3K22 

2,300-  2,725 

2,700-  3,300 
3,300-  4,000 

3,000 

3,000 
3,000 

0  —  200 

0  —  200 
0  —  200 

6.3 

6.3 
6.3 

20w 

20w 
20w 

425  Me 

600  Me 
700  Me 

10  Me 

10  Me 
10  Me 

10-14  db  gain 

Table  1 — Continued.    Sperry  Gyroscope  Special-purpose  Tubes 


2K35 

2  730-  3  330 

3  000 

0  —  200 

6.3 

25w 

600  Me 

3-cavity,  2-stage  cas- 

2K34 
2K47 

2,730-  3,330 
/    250-      280  \ 

3,000 
1  000 

0  —  200 
0  —  200 

6.3 
6  3 

16w 
125  mw 

600  Me 
Input  30  Me 

cade  amplifier,  30  to 
33  db  gain 
3-cavity  oscillator 
buffer  amplifier 
2-cavity  frequency 

-2K46  " 

V2.250-  3,360  J 
(I  730-  3  330  \ 

1,500 

0  —  200 

6.3 

10-70mw 

Output  110  Me 
Input  600  Me 

multiplier 
3-cavity  amplifier-fre- 

V.8, 190-10,000^ 

Output  1,810 
Me 

quency  multiplier 

GASEOUS  CONDUCTION  TUBES 

By  D,  S.  Peck 

For  definitions  of  gas  tube,  anode,  cathode,  etc.,  see  pp.  4-03  to  4-06. 

Arc.  An  arc  is  a  discharge  of  electricity  through  a  gas,  characterized  by  a  change  in 
space  potential  in  the  immediate  vicinity  of  the  cathode  which  is  approximately  equal  to 
the  ionizing  potential  of  the  gas.  (Proposed  for  Standards  for  Pool-cathode  Mercury  Arc 
Power  Converters,  AIEE.) 


13.  GASEOUS  CONDUCTION 

Gaseous  discharges  may  be  classified  in  two  groups  according  to  the  mechanisms  for 
producing  ionization.  The  first  group,  "self-sustaining  discharges,"  includes  those  in 
which  the  energy  for  maintaining  the  discharge  is  supplied  directly  by  the  discharge. 
The  other  group  includes  those  that  require  some  auxiliary  power  in  addition  to  the  energy 
of  the  discharge  itself. 

SELF-SUSTAINING  DISCHARGES.  One  of  the  earliest  known  gaseous-discharge 
devices,  the  Crookes  tube,  is  a  typical  example  of  a  self-sustaining  discharge.  In  that 
tube,  cylindrical  in  form  and  containing  a  low  gas  pressure,  a  luminous  discharge  takes 


GASEOUS  CONDUCTION 


4-59 


place  if  potential  of  sufficient  value  is  applied  between  two  electrodes.  The  appearance 
of  the  glow  is  shown  diagrammatically  with  the  corresponding  voltage  distribution  in 
Fig.  I- 

It  will  be  observed  that  immediately  adjacent  to  the  cathode  is  the  Crookes,  or  cathode, 
dark  space.  Across  this  space  a  large  part  of  the  total  tube  drop  is  concentrated.  This 
drop  is  such  that  ions  moving  toward 
the  cathode  obtain  sufficient  energy  to 
remove  electrons  from  the  cathode  by 
bombardment  and  hence  maintain  ioni- 
zation in  the  tube. 

Following  the  cathode  dark  space  is  a 
luminous  region,  called  the  negative 
glow,  in  which  some  of  the  excited 
atoms  are  returning  to  normal,  radiat- 
ing energy  in  the  form  of  light. 

Following  the  negative  glow  is  an- 
other dark  space  called  the  Faraday 
dark  space,  then  another  luminous  por- 
tion called  the  positive  column  which 
extends  to  the  anode. 

This  general  type  of  discharge  is  used 
for  the  production  of  light  in  many  neon 
and  argon  signs.  It  is  also  employed  in 
protective  tubes  and  in  glow  tubes  used 
for  voltage  regulation,  as  well  as  in  cold- 
cathode  relay  tubes. 

Another  form  of  self-sustaining  dis- 
charge is  found  in  tubes  employing  a 
pool  cathode.  Here  the  current  densi- 
ties run  much  higher  than  in  a  glow 
discharge,  and  the  cathode  dark  space 
becomes  very  small  so  that  extremely 
high  gradients  are  present  close  to  the 
emitting  "spot."  It  is  thus  possible  to 
release  electrons  without  the  high  drop 
generally  found  in  so-called  glow  dis- 
charge tubes  similar  to  the  Crookes 
tube.  With  a  pool  cathode,  emission 


Distance 


TIG.  1.     Voltage  and  Glow  Diistrbution  in  Glow  Dis- 
charge 


appears  to  be  true  field  emission  caused  by  voltage  gradient,  and  not  thermal  emission 
caused  by  the  high  temperature  of  the  spot. 

DISCHARGES  REQUIRING  AUXILIARY  ENERGY.  One  of  the  commonest  forms 
of  this  type  of  discharge  is  the  hot-cathode  tube.  Here  a  heated  filament  or  cathode,  by 
virtue  of  the  thermal  energy  imparted  to  the  surface  molecules,  is  able  to  release  electrons. 
These  electrons  are  drawn  from  the  cathode  by  the  anode  field  and  after  sufficient  travel 
acquire  enough  energy  to  ionize  the  gas  atoms  by  collision.  The  discharge  is  very  similar 
in  structure  to  the  glow  discharge  from  a  cold  cathode,  except  that  the  region  of  cathode 
fall  is  very  much  smaller  and  has  a  lower  voltage  drop  since  the  electrons  are  actually 
ejected  by  the  emitting  properties  of  the  cathode. 

Other  sources  of  ionization  are  possible,  such  as  heat,  photoelectrons,  and  radiation, 

CONTROL  OF  THE  DISCHARGE.  If,  in  a  simple  tube  with  a  cathode  and  an  anode, 
a  third  element  known  as  a  grid  is  introduced  between  cathode  and  anode,  it  is  possible  to 
control  the  starting  of  the  discharge.  If  the  grid  is  made  sufficiently  negative  with  respect 
to  the  cathode,  it  produces  a  retarding  field  at  the  cathode  in  spite  of  the  positive  anode 
potential  and  most  of  the  electrons  emitted  from  the  cathode  are  turned  back  without 
receiving  sufficient  energy  to  cause  ionization.  If  the  grid  is  then  gradually  made  less 
negative,  a  critical  voltage  is  reached  where  some  electrons  acquire  enough  velocity  to 
ionize  the  gas.  Within  a  few  microseconds  the  ionization  builds  up  until  the  current  is 
limited  only  by  the  impedance  of  the  external  circuit. 

Once  ionization  is  complete,  further  changes  in  grid  voltage  have  little  effect  on  the 
discharge  in  most  practical  cases.  If  the  grid  is  made  negative,  positive  ions  from  the  dis- 
charge will  move  toward  it,  causing  a  grid  current  to  flow  and  blanketing  the  grid  suffi- 
ciently to  prevent  its  having  any  further  effect  on  the  arc.  The  grid  becomes  "sheathed" 
with  positive  ions,  the  thickness  of  the  sheath  depending  upon  the  density  of  ionization 
and,  to  some  extent,  upon  the  grid  voltage.  The  sheath  thickness  may  be  of  the  order  of 
hundredths  or  thousandths  of  a  centimeter  under  normal  operating  conditions.  Only  by 


4-60 


ELECTRON  TUBES 


using  grids  with  very  small  openings  and  relatively  large  negative  voltages  is  it  possible 
to  make  the  sheaths  large  enough  to  overlap  and  extinguish  the  discharge.  If,  however, 
the  discharge  ceases,  as  it  would  when  alternating  potential  is  used  for  the  anode  supply, 
the  ionization  diminishes  to  a  low  value  or  disappears  in  a  relatively  short  period  of  time, 
and  the  grid  may  again  exercise  its  control  function. 

VACUUM-TUBE  CONTROL  VS.  GAS-DISCHARGE  CONTROL.  The  character- 
istic of  a  grid-controlled  gaseous-discharge  tube  of  passing  either  no  current  or  full  current, 
together  with  the  normal  inability  of  the  grid  to  stop  the  discharge,  are  two  important 
differences  from  hard-vacuum  grid-controlled  tubes.  Although  both  types  possess  the 
unidirectional  conduction  properties  of  a  rectifier,  the  hard-vacuum  tube  with  its  con- 
tinuous control  may  be  likened  to  a  rheostat  in  series  with  a  circuit,  whereas  the  gas-filled 
control  tube  may  be  considered  similar  to  a  switch  which  can  be  closed  at  any  time  but 
opened  only  when  current  is  not  flowing.  Obviously,  the  hard-vacuum  tube,  which  func- 
tions like  a  rheostat,  must  be  capable  of  dissipating  the  power  losses  caused  by  pasage  of 
current  through  the  tube  when  the  voltage  across  the  tube  is  increased.  The  low  voltage 
drop  (from  about  5  to  15  volts  in  most  hot-cathode  tubes)  existing  when  the  gas  tube  is 
passing  current  results  in  relatively  low  losses  in  the  tube,  and  there  is  no  plate  dissipation 
whatever  when  the  tube  is  turned  off.  Thus,  a  gas  tube  is  capable  of  handling  much 
heavier  currents  and  more  power  than  a  hard-vacuum  tube  of  the  same  size. 

Because  of  the  difference  in  control  characteristic  the  circuit  technique  used  with  gas 
tubes  is  entirely  different  from  that  employed  with  hard-vacuum  tubes.  For  instance, 
the  gas  tube  may  be  used  as  a  sensitive  relay  to  operate  a  contactor  when  the  grid  voltage 
of  the  tube  reaches  a  given  value.  With  a  voltage  of  adjustable  phase  relation  on  the 
grids  it  may  be  used  as  a  controlled  rectifier  tube  to  control  average  output  voltage.  The 
gas  tube  may  also  be  used  in  suitable  circuits  to  change  direct  current  to  alternating  cur- 
rent or  alternating  current  of  one  frequency  to  alternating  current  of  a  different  frequency. 


14.  THYRATRON  TUBES 

A  thyratron  is  a  hot-cathode,  gas-discharge  tube  in  which  one  or  more  electrodes  are 
employed  to  control  electrostatically  the  starting  of  the  unidirectional  current  flow.  (IRE 
Standard.) 

These  types  of  tubes  cover  an  intermediate  power  range.  Tubes  are  available  in  sizes 
up  to  12.5  amp  average  current  and  up  to  15,000  volts  peak.  (See  Table  1.) 

Similar  gas  tubes  are  built  without  grids  for  use  as  rectifiers.    Since  their  major  single 
application  is  in  transmitting  circuits,  they  are  listed  in  article  13  with  transmitter  tubes. 
CONSTRUCTION.     Figure  2  shows  a  typical  construction  of  a  simple  filamentary- 
type  gas  tube.    The  filament  is  in  the  form  of  a  ribbon  of  nickel  or  nickel-cobalt  alloy, 

formed  in  a  helical  or  S-shaped  form,  and 
coated  with  an  electron-emissive  coating.  A 
grid  in  the  form  of  a  cylinder  is  mounted  as 
shown,  and  supported  by  a  clamp  from  the 
lower  stem.  The  grid  may  also  be  mounted 
on  the  stem  leads  and  supported  additionally 
from  the  glass  at  the  top  of  the  tube.  The 
nickel  anode  in  the  form  of  a  cup  is  mounted 
by  means  of  another  glass  stem  at  the  top.  A 
washerlike  cross  piece  in  the  grid  determines 
the  control  characteristic  by  its  relative  spac- 
ing between  cathode  and  anode  and  by  the 
size  and  shape  of  a  hole  in  the  washer.  The 
sides  of  the  grid  shield  the  control  area  from 
effects  of  charges  collected  on  the  glass  walls 
and  from  extraneous  fields.  Carbonized  nickel 
is  frequently  used  for  these  parts  because  of  its 
heat-radiating  properties,  but  bright  nickel  is 
PIG.  3.  Shield-grid  Type  sometimes  necessary  in  inert-gas-filled  tubes. 


FIG.    2.      Filamen- 
tary-type      Thyra- 
tron Tube 


of  Thyratron  Tube '      Argon,  xenon,  and  mercury  vapor  are  the 

most  common  mediums  for  gas  tubes. 
FOUR-ELECTRODE  CONSTRUCTION.  Figure  3  shows  a  sketch  of  a  shield-grid 
type  of  construction.  The  large  shield  grid,  generally  held  at  a  fixed  potential,  permits 
the  use  of  a  small  control  grid  of  extremely  high  effective  input  resistance.  The  shield 
grid  shields  the  control  grid  from  the  heat  of  the  cathode  and  anode  and  minimizes  the 
possibility  of  sputtered  or  evaporated  material  from  the  cathode  contaminating  the  grid, 


THYKATRON  TUBES 


4-61 


Table  1.    Available  Types  of  Thyratrons 


Type 

Designation 

Anode  Current 

Peak  Anode 
Voltage 

Cathode 
/  =  filament 
k  =  heater 

loniz- 
able 
Me- 
dium 

Remarks 

Aver- 
age 

Peak 

Averag- 
ing 
Time, 
seconds 

For- 
ward 

In- 
verse 

Volt- 
age 

Cur- 
rent 

2C4 

S 

0.005 

0.020 

30 

450 

450 

2.5k 

0.65 

Gas 

Negative  control  tube 

297A 

WE 

0.010 

0.060 

250 

250 

1.75/ 

0.35 

Gas 

>Jpcr<}-f-?VP   f»nnfpn1    fnKp 

546 

GL- 

0.020 

0.100 

15 

500 

500 

6.3k 

0.15 

Gas 

Shield  grid  tube 

269A 

WE 

0.020 

0.20 

275 

275 

2.2f 

0.55 

Gas 

6D4 

S 

0.025 

0.100 

30 

450 

450 

Gas 

Miniature  negative 

control  tube 

233A 

RX- 

0.025 

1.5 

,500 

1,500 

2.5f 

2.5 

Gas 

"NTpo^itivp       nf     If  iKp 

629 

WL- 

0.040 

0.2 

10 

350 

350 

2.5k 

Gas 

Negative  control  tube 

2051 

Ray,  Ch, 

0.075 

0.375 

30 

350 

700 

6.3h 

0^6 

Gas 

Shield  grid  tube 

GL-,  RCA 

884 

RX-,  Ch,  S, 

0.075 

0.30 

30 

350 

350 

6.3h 

0.6 

Gas 

Sweep-circuit  tube 

WL-, 

RCA,  GL- 

885 

RX-,  Ch,  S, 

0.075 

0.30 

30 

350 

350 

2.5h 

1.5 

Gas 

Sweep-circuit  tube 

RCA, 

GL-,  WL- 

610 

KU- 

0.10 

0.40 

10 

500 

500 

2.5f 

6.5 

Gas 

Positive  control  tube 

636 

KU- 

0.10 

0.40 

15 

350 

350 

2.5f 

7.5 

Gas 

Negative  control  tube 

2D21 

RCA 

0.10 

0.50 

30 

650 

1,300 

6.3A 

0.6 

Gas 

Shield-grid  tube 

338A 

WE 

0.10 

0.60 

325 

325 

10.  Oh 

0.50 

Gas 

Negative  control  tube 

2050 

Ray,  GL-, 

0.10 

1.00 

30 

650 

1,300 

0.60 

Gas 

Shield-grid  tube 

Ch,  WL-, 

RCA 

502A 

GL-.WL- 

0.10 

1.0 

30 

650 

1,300 

6.3A 

0.60 

Gas 

Shield-grid  tube 

2A4G 

S,  Ch,  Ray 

0.10 

1.25 

45 

200 

200 

2.5/ 

2.5 

Gas 

Negative  control  tube 

178  A 

FG- 

0.125 

0.50 

15 

500 

500 

2.5/ 

2.25 

Gas 

Negative  control  tube 

17 

FG-,  WL- 

0.50 

2.0 

15 

2,500 

5,000 

2.5/ 

5.0 

Hg 

Negative  control  tube 

967 

UE- 

0.50 

2.0 

15 

2,500 

2,500 

2.5/ 

5.0 

Hg 

Negative  control  tube 

81A 

FG-,  WL- 

0.50 

2.0 

15 

500 

500 

2.5/ 

5.0 

Gas 

Negative  control  tube 

98A 

FG- 

0.50 

2.0 

15 

500 

500 

2.5/ 

5.0 

Gas 

Shield-grid  tube 

97 

FG- 

0.50 

2.0 

15 

1,000 

1,000 

2.5/ 

5.0 

Hg 

Shield-grid  tube 

627 

WL-,  GL- 

0.64 

2.5 

30 

1,250 

2,500 

2.5/ 

6.0 

Hg 

Negative  control  tube 

394A 

WE 

0.64 

2.5 

5     ' 

1,250 

1,250 

2.5/ 

3.25 

Hg  and 

Negative  control  tube 

gas 

3D22 

RCA 

0.75 

6.0 

30 

650 

1,300 

6.3A 

2.6 

Gas 

Shield-grid  tube 

GIB 

EL- 

.0 

8.0 

450 

700 

2.5/ 

6.3 

Gas 

Negative  control  tube 

303 

CE- 

.0 

8.0 

450 

700 

2.5/ 

6.0 

Gas 

Negative  control  tube 

302 

CE- 

4^5 

1,000 

1,000 

2.5/ 

7.0 

Hg 

Negative  control  tube 

287A 

WE 

.5 

6.0 

5 

500 

500 

2.5/ 

7.0 

Hg 

Negative  control  tube 

323A 

WE 

.5 

6.0 

5 

500 

500 

2.5/ 

7.0 

Hgand 

Negative  control  tube 

gas 

393A 

GL-,  WE 

1.5 

6.0 

5 

1,250 

1,250 

2.5/ 

7.0 

Hgand 

Negative  control  tube 

gas 

3C23 

GL-.WL- 

1.5 

6.0 

5 

1,250 

1,250 

2-5/ 

7.0 

Hg  and 

Negative  control  tube 

gas 

678 

WL-,  GL- 

1.6 

6.0 

1  cycle 

15,000 

15,000 

5.0A 

7.5 

Hg 

Negative  control  tube 

21 

KY- 

3.0 

11,000 

2.5/ 

10.0 

Hg 

Transmitter  keying 

tube 

628 

KTJ- 

2.0 

8.0 

30 

1,250 

2,500 

5.0/ 

11.5 

Hg 

Negative  control  tube 

305 

CE- 

2.0 

12.0 

850 

,700 

2.5/ 

6.5 

Gas 

Negative  control  tube 

27A 

FG- 

2.5 

10.0 

15 

1,000 

,000 

5.0/ 

4.5 

Hg 

Negative  control  tube 

973 

UE- 

2.5 

10.0 

15 

3,000 

,000 

5-0/ 

6.75 

Hg 

Negative  control  tube 

154 

FG- 

2.5 

10.0 

15 

500 

500 

5.0/ 

7.0 

Gas 

Shield-grid  tube 

33 

FG-,  WL- 

2.5 

15.0 

15 

,000 

,000 

5.0A 

4.5 

Hg 

Negative  control  tube 

57 

FG-,  WL- 

2.5 

15.0 

15 

,000 

,000 

5.0h 

4.5 

Hg 

Negative  control  tube 

67 

FG- 

2.5 

15.0 

15 

,000 

,000 

5.0h 

4.5 

Hg 

Inverter  tube 

95 

FG- 

2.5 

15.0 

15 

,000 

,000 

5.0k 

4.5 

Hg 

Shield-grid  tube 

672 

WL-.GL- 

2.5 

30.0 

15 

,500 

,500 

5.0k 

6.0 

Hg 

Shield-grid  tube 

C3J 

EL- 

2.5 

30.0 

750 

,250 

2.5f 

9.0 

Gas 

Negative  control  tube 

632-A 

WL- 

2.5 

30.0 

15 

1,500 

,500 

5.0k 

6.0 

Hg 

Shield-grid  tube 

677 

WL- 

4.0 

15.0 

15 

10,000 

10,000 

5.0k 

10.0 

Hg 

Negative  control  tube 

354A 

WE 

4.0 

16.0 

15 

1,500 

1,500 

2.5f 

16.0 

Hg 

Negative  control  tube 

355A 

WE 

4.0  ' 

16.0 

15 

350 

350 

2.5f 

16.0 

Hgand 

Negative  control  tube 

gas 

4-62 


ELECTRON  TUBES 


Table  1.    Available  Types  of  Thyratrons — Continued 


Type 

Designation 

Anode  Current 

Peak  Anode 
Voltage 

Cathode 
/  =  filament 
h  =  heater 

loniz- 
able 
Me- 
dium 

Remarks 

Aver- 
age 

Peak 

Averag- 
ing 
Time, 
seconds 

For- 
ward 

In- 
verse 

Volt- 
age 

Cur- 
rent 

105 

172 
676 
C6J 
306 
C6C 
624 
C16J 
41 
414 

Ch,  WI^, 
FO- 
WL-, FG- 
KTJ- 
EL- 
CE- 
EL- 

Wlr 

EL- 
WL-,  FG- 
GL-.WL- 

6.4 

6.4 
6.4 
6.4 
6.4 
6.4 
6.4 
12 
12.5 
12.5 

40.0 

40.0 
40.0 
77.0 
77.  0 
77.  0 
77.0 
100.0 
75.0 
100.0 

15 

15 
15 

2,500 

2,000 
2,500 
750 
750 
2,000 
2,500 
750 
10,000 
2,000 

2,500 

2,000 
2,500 
1,250 
1,250 
4,000 
2,500 
1,250 
10,000 
2,000 

5.0/i 

5.0A 
5.0A 
2.5/ 
2.5/ 
2.5/ 
5.0/i 
2.5/ 
5.0/1 
5.0A 

10.0 

10.0 
10.0 
21.0 
18.0 
24.0 
10.0 
31.0 
20.0 
20.0 

Hg 

Hg 
Hg 
Gas 
Gas 
Gas 
Hg 
Gas 
Hg 
Hg 

Shield-grid  tube 

Shield-grid  tube 
Negative  control  tube 
Negative  control  tube 
Negative  control  tube 
Negative  control  tube 
Negative  control  tube 
Negative  control  tube 
Inverter  tube 
Metal   negative   con- 
trol tube 

15 

30 
30 

NOTES 


Prefix  Used  ly 

GL-  General  Electric  Company 

FG-  General  Electric  Company 

WL-  Westinghouse  Electric  Corporation 

KU-  Westinghouse  Electric  Corporation 

KY-  Eimac 

TIE-  United  Electronics 

EL-  Electrons,  Inc. 


Prefix  Used  ly 

CE-  Continental  Electric  Company 

RX-  Raytheon 

Letter  Indicates 

S  Sylvania  Electric  Products,  Inc. 

Ch  Chatham  Electronics 

Ray  Raytheon 

WE  Western  Electric  Company 

RCA  Radio  Corporation  of  America 


•with  consequent  reduction  in  grid  emission.  The  shield  grid  may  act  also  as  an  electro- 
static shield  to  prevent  sudden  voltage  fluctuations  of  the  anode  from  inducing  transient 
voltages  on  the  grid  with  consequent  loss  of  control.  Of  course,  it  is  also  possible  to  use 
tubes  of  this  shield-grid  type  with  "signal'*  voltages  on  both  electrodes  so  that  operation 
of  the  tube  is  a  function  of  both  grid  potentials. 

METAL  TUBES.  Metal  envelopes  are  used  for  many  industrial  tubes,  particularly  in 
the  larger  sizes,  because  of  the  greater  sturdiness  and  dependability  of  metal  than  of  glass 
and  to  facilitate  mounting  the  tubes  on  a  panel.  In  the  medium  or  smaller  sizes  this  con- 
struction frequently  has  no  real  advantage  over  glass  tubes  because  internal  tube  elements 
may  be  as  subject  to  breakage  as  the  envelope  under  conditions  of  shock  or  vibration. 


15.  VOLTAGE  LIMITS  OF  THYRATRONS 

PEAK  INVERSE  ANODE  VOLTAGE.  Peak  inverse  anode  voltage  is  the  maximum 
instantaneous  anode  voltage  in  the  direction  opposite  to  that  in  which  the  tube  is  designed 
to  pass  current.  (IRE  Standards  on  Electronics,  1938.) 

The  maximum  peak  inverse  voltage  which  can  be  applied  is  a  function  of  the  shape  and 
spacing  of  the  electrodes,  the  current  conducted,  and  the  gas  pressure.  In  thyratrons 
having  argon  or  xenon  or  some  other  inert  gas,  the  arcback  potential  is  relatively  inde- 
pendent of  the  tube  temperature.  In  mercury-vapor  tubes,  however,  the  mercury-vapor 
pressure  doubles  roughly  with  every  10  deg  cent  increase  so  that  the  maximum  peak  in- 
verse voltage  is  seriously  affected.  Figure  4  shows  a  typical  curve  of  arcback  voltage  vs. 
temperature  for  a  mercury-vapor  tube.  In  rating  a  tube  of  this  class,  therefore,  it  is 
necessary  to  specify  not  only  the  peak  inverse  voltage  but  the  maximum  condensed- 
mercury  temperature  as  well. 

In  designing  a  tube  for  high-voltage  operation,  it  is  sometimes  necessary  to  constrict 
completely  the  space  at  the  back,  or  top,  of  the  anode,  so  that  it  is  impossible  for  any 
discharge  to  take  place  between  the  anode  and  cathode  around  the  outside  of  the  grid. 

PEAK  FORWARD  VOLTAGE.  Peak  forward  voltage  is  the  maximum  instantaneous 
anode  voltage  in  the  direction  in  which  the  tube  is  designed  to  pass  current.  (IRE  Stand- 


CURRENT  LIMITS  OF  THTRATRONS 


4-63 


CO 
O 

o 
o 


rs 

Oi 
O 

o 
o 


-4000 


nod 

M 
0 

8 


Itage 


.Tern  i 


ards  on  Electronics,  1938.)  The  same 
factors  of  tube  geometry  and  current 
affect  the  maximum  permissible  peak 
forward  voltage.  In  addition  the  grid 
must  be  so  designed  as  to  maintain 
the  proper  characteristics  up  to  the 
voltage  desired. 

TESTS.  The  peak  forward  voltage 
may  be  tested  by  measuring  the  con- 
trol characteristic  at  the  maximum 
temperature  ratings,  although  a  com- 
mon test  is  a  full-load  operation  of  the 
tubes  in  a  circuit  giving  both  peak 
inverse  and  peak  forward  voltages. 
This  operation  continues  at  the  rated 
average  current  of  the  tubes  for  a 

length  of  time  sufficient  to  stabilize 

the  tube  temperature,  at  which  time  v       40  50  60  70 

the  grid  control  is  checked.     Thejfre--,      ,     ^  ,..        ®°nderise^8  Temperature 
quency  of  arcback  is  also  observed.    3^££^ 
The  severity  of  the  test  varies  con- 
siderably with  the  current  and  voltage  conditions  at  the  time  of  current  commutation 
from  one  tube  to  the  next;  therefore  the  type  of  operation  circuit  used  is  important. 
When  simple  circuits  are  used  the  voltage  is  frequently  set  above  the  rating  to  make  the 
test  sufficiently  severe. 


16.  CURRENT  LIMITS  OP  THYRATRONS 

Most  commercial  tubes  have  cathodes  made  by  coating  a  base  metal  such  as  nickel  or  a 
nickel  alloy  with  one  of  two  types  of  coatings.  The  first  is  barium  oxide  or  a  mixture  of 
barium,  strontium,  and  calcium  oxides,  formed  by  coating  with  the  carbonates  and  reduc- 
ing to  the  oxides  by  high  temperature  during  the  processing.  The  second  is  a  molecular 
mixture  of  barium  oxide  and  nickel  oxide  known  as  barium  nickelate,  formed  by  coating 
with  barium  carbonate  and  a  nickel  oxide  before  processing. 

Cathode  construction  is  of  two  general  types,  the  filamentary  and  the  indirectly  heated. 
The  filamentary  type  is  simple  in  construction  and  has  a  relatively  short  heating  time. 
The  usual  absence  of  heat  shielding,  however,  reduces  the  efficiency,  requiring  more  watts 
per  ampere  of  emission,  and  increases  the  heat  that  the  tube  must  dissipate.  The  in- 
directly heated  type  inherently  lends  itself  to  more  efficient  use  of  the  cathode  surface, 
thereby  reducing  the  heating  power  required.  It  has  the  disadvantage  of  greater  expense 
and  longer  heating  time,  a  factor  that  is  becoming  more  and  more  important. 

By  raising  the  cathode  temperature  it  is  usually  possible  to  raise  the  emission  per  unit 
area  and  the  emission  per  watt  heating  power.  However,  this  raises  the  rate  of  coating 
evaporation  and,  if  carried  too  far,  shortens  the  life  of  the  tube. 

PEAK  ANODE  CURRENT.  The  peak  current  that  a  tube  is  capable  of  passing  with- 
out harm  depends  upon  the  area,  the  temperature,  and  the  geometry  of  the  cathode 
surface,  and  somewhat  upon  the  pressure  of  the  gas.  If  the  rated  value  is  exceeded,  the 
tube  voltage  drop  may  increase  so  that  destructive  ion  bombardment  of  the  catbode 
occurs.  Also,  particles  of  coating  may  be  mechanically  sputtered  from  the  cathode  to  the 
grid,  where  they  are  a  potential  source  of  grid  emission.  There  is  the  further  possibility 
of  sudden  cessation  of  current  flow,  particularly  at  low  gas  pressure  where  the  ion  density 
is  insufficient  to  neutralize  the  space  charge  at  points  where  the  discharge  is  the  most 
dense.  This  is  commonly  known  as  "starvation"  and  may  result  in  voltage  surges  in  the 
circuit  in  which  current  has  been  flowing,  sometimes  breaking  down  insulation  of  the 
circuit  components.  The  barium  nickelate  coating  is  less  subject  to  this  trouble  since  the 
coating  is  more  conductive  and  will  stand  a  higher  arc  drop  without  sputtering  or  surging. 

AVERAGE  ANODE  CURRENT.  Because  of  the  essentially  constant  tube  voltage 
drop  over  the  current  range,  tube  heating  during  operation  is  a  function  of  average  current, 
rather  than  rms  current  as  in  most  electrical  apparatus.  Overloading  causes  excessive 
heating  of  all  the  tube  electrodes  and  may  result  in  failure  because  of  (1)  evolution  of 
sufficient  foreign  gas  to  render  the  tube  inoperative,  (2)  grid  emission  and  loss  of  control 
due  to  high  grid  temperature,  (3)  arcback  due  to  the  high  temperature  of  the  anode,  (4) 
short  cathode  life  due  to  high  evaporation,  or  (5)  mechanical  failure  of  the  envelope  or 
seals. 


4-64 


ELECTRON  TUBES 


ANODE  CURRENT  AVERAGING  TIME.  When  the  load  is  fluctuating,  as  it  might 
with  a  motor  load  or  welding  control,  the  average  current  must  be  calculated  over  a 
specified  period  of  time  chosen  so  as  to  include  the  worst  current  conditions.  For  example, 
consider  two  tubes  having  a  maximum  average  current  of  12.5  amp,  a  maximum  peak 
current  rating  of  75  amp,  and  an  averaging  time  of  30  sec  in  a  biphase  half-wave  rectifier 
with  sufficient  inductance  to  make  the  d-c  ripple  negligible.  These  tubes  could  supply  to 
the  load  75  amp  for  10  sec,  50  amp  for  15  sec,  or  25  amp  for  30  sec  out  of  every  30  sec 
without  exceeding  either  peak  or  average  rating. 

GRID  CURRENTS.  In  order  to  prevent  overloading  of  the  cathode  or  the  grid  through 
the  grid  circuit,  a  maximum  instantaneous  grid  current  and  a  maximum  average  grid  cur- 
rent rating  are  generally  given.  Under  most  conditions  of  operation,  grid  currents  are 
much  less  than  these  ratings. 

ANODE  SURGE  CURRENT.  The  anode  surge  current  is  the  current  which  would  be 
conducted  through  the  anode  under  fault  conditions.  The  maximum  surge  current  rating 
is  a  measure  of  the  ability  of  the  tube  to  withstand  extremely  high  transient  currents. 
The  tube  should  carry  the  specified  current  for  not  longer  than  a  given  length  of  time  in 
the  event  of  short  circuit,  but  it  should  not  be  expected  to  carry  repeated  short  circuits 
without  a  reduction  of  life  and  the  possibility  of  immediate  failure.  This  rating  forms  a 
basis  for  set  design  to  obtain  best  tube  performance.  If  sufficient  impedance  is  present  to 
limit  the  fault  current  to  this  rating,  not  only  will  the  tube  be  able  to  carry  that  current 
in  the  event  of  a  fault  in  the  circuit,  but  also  the  possibility  of  a  fault  in  the  tube  itself 
seems  to  be  reduced. 

TESTS.  Tests  for  maximum  average  current  are  generally  made  by  operating  the  tube 
at  full  load  and  checking  the  other  ratings  which  are  dependent  upon  tube  temperature, 
as  described  in  other  sections. 

As  has  been  stated,  the  peak  current  rating  is  a  function  of  the  emissive  capabilities  of 
the  cathode,  and  the  best  test  of  this  is,  after  all,  satisfactory  life  while  operating  at  that 
peak  current.  Several  methods  have  been  used  for  testing  the  emission,  however.  A  simple 
test  is  to  conduct  an  average  current  through  the  tube  of  a  value  between  the  average 
and  the  peak  ratings  and  then  measure  the  tube  voltage  drop  with  a  d-c  meter  across  the 
tube  or  with  the  deflecting  plates  of  an  oscilloscope  connected  from  anode  to  cathode. 
This  method  does  not  discriminate  very  well  between  good  tubes  and  bad  except  for  some 
small  tubes  where  the  average  current  may  be  held  fairly  high  for  the  test.  It  will  suffice 
for  a  rough  check  on  any  tube,  if  used  within  the  rating. 

If,  with  the  oscilloscope  and  a  suitable  d-c  amplifier  across  the  tube,  the  tube  is  allowed 
to  conduct  the  peak  rated  current  for  only  a  few  half-cycles  each  second,  the  cathode 

temperature  is  not  altered  appreci- 
ably by  the  load  current,  and  the 
tube  drop  may  be  used  as  an  accu- 
rate indication  of  cathode  quality. 
Figure  5  shows  a  typical  trace  ob- 
served on  a  good  tube,  and  Fig.  6 
shows  a  low-emission  tube  in  which 
the  drop  rises  to  an  excessive  value 
at  the  maximum  current  point. 

Another  method  of  testing  the 
emission  is  to  conduct  half-cycle 
pulses  as  above,  but  of  increasing 
current  magnitude,  so  that  the  point 
is  finally  reached  where  the  cathode 
"sparks"  or  sputters,  giving  a  broken  voltage  drop  trace.  The  "sparking  point"  can  be 
calibrated  against  the  peak  current  rating  to  give  a  proper  test. 

In  all  such  tests,  the  temperature  of  mercury-vapor  tubes  should  be  controlled  quite 
accurately  by  an  oil  bath  or  controlled-flow  air  bath. 

Surge  current  tests  are  generally  made  by  the  manufacturer  on  a  given  design  of  tube 
to  insure  a  construction  sufficiently  rugged  for  ordinary  service.  Since  these  tests  detract 
from  the  life  of  the  tube,  they  are  not  part  of  the  normal  test  procedure.  In  general,  the 
test  is  made  by  passing  the  rated  surge  current  through  the  tube  with  some  protective 
device  or  control  device  arranged  to  open  in  a  definite  time,  generally  0.1  sec.  After  the 
tube  has  been  subjected  to  one  or  more  overloads  of  this  type,  it  is  given  an  operation 
test  for  general  performance. 


FIG.  5.  t  Trace  on  Cathode- 
ray  Oscilloscope  Indicating 
Good  Emission 


FIG.  6.    Trace  on  Cathode- 
ray  Oscilloscope  Indicating 
Poor  Emission 


CONTROL  CHARACTERISTICS 


4-65 


500 


17.  CONTROL  CHARACTERISTICS 

Control  Characteristic.  The  control  characteristic  of  a  gas  tube  is  a  relation,  usually 
shown  by  a  graph,  between  critical  grid  voltage  and  anode  voltage.  (IRE  Standards  on 
Electronics,  1938.) 

Critical  Grid  Voltage.     Critical  grid  voltage  in  a  gas  tube  is  the  instantaneous  value  of 
grid  voltage  when  the  anode  current  starts  to 
flow.     (IRE  Standards  on  Electronics,  1938.) 

Critical  Anode  Voltage.  The  critical  anode 
voltage  of  a  gas  tube  is  the  instantaneous 
anode  voltage  when  the  anode  current  starts 
to  flow. 

As  previously  explained,  once  a  thyratron 
is  passing  anode  current  the  grid  has  little 
effect  on  the  anode  current,  which  is  then 
limited  only  by  the  impedance  of  the  load  in 
the  anode  circuit.  There  are,  therefore,  no 
thyratron  characteristic  curves  relating  anode 
potential,  grid  potential,  and  anode  current 
as  there  are  for  vacuum  tubes.  The  relation- 
ship between  anode  voltage  and  grid  voltage 
which  just  permits  conduction,  known  as  the 
control  characteristic,  is  of  considerable  im- 
portance, however. 

In  an  inert-gas-filled  tube,  there  is  a  neg- 
ligible change  of  characteristic  with  normal 
temperature  changes.  There  are  initial  vari- 
ations between  tubes,  however,  and  there  are 
further  changes  in  any  given  tube  with  life, 
the  characteristic  shifting  slightly  more  nega- 
tive as  the  emission  becomes  completely 
stable,  and  then  shifting  more  positive  as  the 
end  of  the  tube  life  approaches.  Figure  7 
shows  the  typical  control  characteristic  of  a 
small  thyratron,  the  range  of  curves  covering 
all  variations  between  tubes  and  with  life. 
This  shows  the  equipment  designer  what  range  he  may  expect  and  must  design  for.  In 
mercury-vapor  tubes,  the  characteristic  changes  greatly  with  mercury  temperature  and 
the  temperature  must  be  specified  for  such  a  curve,  or  the  range  must  be  extended  to 


-8     -7    -6     -5     -4     -3     -2      -1 
Grid  voltage,  volts 

FIG.  7.    Typical  Thyratron  Control  Character- 
istic.    (Shaded  Area  Shows  Range  of  Character- 
istics.) 


3600 
3200 
2800 
g.2400 

0 

>  2000 
<  1600 
Q  1200 
800 
400 

•4 

L, 

Sfc 

y 

1 

& 

*b 

1°'' 

V 

s^ 

^s 

\ 

\ 

\ 

\ 

\ 

s^S 

X 

\ 

\ 

\| 

x 

^\ 

X 

\\ 

(c 

on 

der 

se 

d  H 

?  ' 

Te 

m 

n° 

C.) 

X 

s^ 

SX 

ss 

\ 

\ 

X 

S^x 

^\ 

\ 

\ 

\ 

> 

^1 

N\ 

s, 

\ 

\ 

S 

^ 

s\ 

^ 

\ 

\ 

^ 

^ 

s^S 

s, 

^ 

\ 

^ 

s^S 

\ 

S 

s 

N^5 

s^s 

s 

\ 

\ 

N^ 

x 

\ 

^ 

^ 

\ 

X 

\ 

^ 

^ 

\ 

^ 

v. 

\ 

F 

3-1 

7 

^ 

\ 

V 

S 

S{ 

^ 

\ 

\ 

\ 

S 

\ 

X 

x 

^ 

^s 

N 

"v, 

X 

*«*B 

««^ 

•^ 

^- 
— 

:^S 

-14  -13  -J.2   -11  —10    -9     -8     -7     -6     —5     -4     -3      -2      -1 
Grid  Voltage  at  Start  of  Discharge 

FIG.  8.     Control  Characteristics  for  a  Mercury-vapor  Thyratron 

include  the  variations  within  the  published  temperature  limits.    Figure  8  shows  the  vari- 
ation in  the  control  characteristic  of  a  mercury  tube  with  varying  temperature. 


4-66 


ELECTRON  TUBES 


aooo 


1600 


It  will  be  noted  that  the  criticargrid  voltages  of  the  tubes  shown  are  negative  over  the 
greater  part  of  the  operating  range.  As  long  as  the  grid  is  more  negative  than  the  cathode, 
electrons  emitted  from  the  cathode  cannot  flow  to  the  grid,  and  there  is  no  grid  current 
from  this  source.  Control  is  possible  therefore  with  the  use  of  a  very  small  amount  of  grid 
power,  which  is  the  advantage  of  such  a  "negative  grid"  tube. 

"Positive  grid"  tubes  are  available  in  which  it  is  necessary  to  force  the  grid  positive 
over  the  entire  operating  range  to  fire  the  tube.  To  assure  non-conduction  it  is  required 
only  that  the  grid  circuit  be  opened  or  the  grid  held  at  zero  voltage  or  allowed  to  float. 

These  application  advantages 
are  offset,  however,  by  the  fact 
that  considerable  power  must  be 
available  to  drive  this  type  of 
tube,  and  sometimes  a  discharge 
takes  place  between  grid  and 
cathode  before  the  anode-cathode 
path  becomes  ionized. 

SHIELD-GRID  CHARAC- 
TERISTICS. In  four-electrode 
tubes  the  firing  point  becomes  a 
function  of  the  potential  of  both 
grids.  Figure  9  shows  how  the 
control  characteristic  curve 
varies  with  shield-grid  potentials. 
With  negative  shield-grid  poten- 
tials, however,  the  variations 
between  tubes  become  excessive 
with  many  types.  The  shield- 
grid  signal  may  be  used  as  a  con- 
trol to  switch  the  characteristic 
in  or  out  of  the  operating  range 
of  the  circuit,  and  in  some  cases 
may  even  be  used,  with  the  nec- 
essary adjustments,  to  obtain  a 
desired  control  curve. 


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—14    -12     -JO     -8       -6       -4       -2         0      +2       4-4 
Control  Grid  Voltage 

FIG.  9.     Control  Characteristics  for  Shield-grid  Type  of  Thy- 
ratron,  Showing  Effect  of  Shield  Potential 


GRID  CURRENTS.  The  grid  currents  in  a  thyratron,  both  before  and  after  discharge, 
are  extremely  important  in  deterrnining  the  necessary  grid  power  or  the  permissible 
impedance  in  the  grid  circuit. 

Critical  grid  current  in  a  gas  tube  is  the  instantaneous  value  of  grid  current  when  the 
anode  current  starts  to  flow.  If  a  high  grid  resistance  is  used  and  this  critical  grid  current 


—22   -20   -18  -16  -14  -12  —10   —8     -6    -4     -2 


FIG.  10.     Grid  Voltage-Grid  Current  Curves  after  Breakdown  in  Three-electrode  Thyratron 


becomes  excessive,  owing  to  overheating  and  grid  emission,  enough  voltage  drop  across 
the  grid  resistance  may  result  so  that  the  available  grid  supply  voltage  is  not  sufficient  to 


CONTROL  CHARACTERISTICS 


4-67 


800 


600 


>400 


U- 


Tub's  ( 


control  the  tube.    This  current,  then,  becomes  a  limitation  on  the  grid  supply  voltage  and 
impedance  and  on  the  time  constant  with  which  the  grid  circuit  can  operate. 

Figure  10  shows  the  grid  current  in 
a  2.5-amp  average,  three-electrode 
thyratron  after  discharge  has  occurred. 
Negative  values  indicate  positive  ion 
current  to  the  grid  from  the  discharge, 
and  positive  values  indicate  electron 
current.  A  four-electrode  tube,  other- 
wise similarly  designed,  would  have 
roughly  one-tenth  the  amount  of  grid 
current  under  these  conditions. 

Figure  11  shows  the  grid  current  for 
various  anode  voltages  immediately 
before  the  start  of  discharge  on  the 
same  three-electrode  tube,  and  Fig.  12 
shows  the  same  for  the  equivalent 
four-electrode  tube. 

TESTS.  The  control  characteristic 
should  be  tested  to  the  specified  limits, 


FIG.   11. 


Grid  Current  before   Breakdown  in  Three- 
electrode  Thyratron 


-2  -1 

Grid  Ctrrreat  in  Microamperes 
usually  at  two  or  more  values  of  anode 
voltage.  A  sufficient  d-c  control  grid 
voltage  should  be  applied  to  prevent 
conduction  (or  firing) ,  through  a  low  grid  resistance.  The  specified  d-c  anode  voltage  is 
applied  and  the  control  grid  gradually  made  more  positive  until  breakdown  occurs  to 
the  anode,  at  which  time  the  critical  grid  voltage  is  observed.  Condensed-mercury 

temperature  should  be  controlled. 
The  critical  anode  voltage  may 
be  observed  by  holding  the  grid  at 
zero  voltage  and  increasing  the 
anode  supply  until  firing  occurs. 

The  critical  grid  current  is  gen- 
erally measured  as  shown  in  Fig. 
13.  The  tube  is  first  operated  at 
the  full  average  current  rating  as 
indicated  by  a  d-c  ammeter  in  the 
anode  circuit,  in  order  to  heat  all 
the  parts  to  their  operating  tem- 
peratures. With  the  grid  resistor 
rg  short-circuited,  the  grid  supply 
voltage  is  then  made  more  nega- 
tive until  the  tube  ceases  to  con- 
duct. This  is  possible  since  an  a-c 
supply  is  used  and  the  tube  does 
not  conduct  during  the  negative 
half-cycle.  This  voltage  reading  is 
denoted  by  V\.  Another  reading 
is.  ^^dmtely  taken  with  rg  in  the 
circuit.  With  most  tubes,  a  value 


800 

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JOS           0.004           0.003            0.002           0.001               C 

Grid  Current  in  Microamperes 


FIG.  12.    GridT  Current  before  Breakdown  in  Shield-grid  Thy- 
ratron Tube 

of  rg  between  10  and  100  megohms 

is  sufficient  to  make  the  second  reading,  Vz,  considerably  higher  than  Vi.  Since  the  actual 
critical  grid  voltage  measured  directly  at  the  grid  is  the  same  in  both  measurements,  the 
difference  in  the  two  readings  must  be  accounted  for  as  voltage  drop  in  the  resistor  r&. 
The  grid  current,  therefore, 
is  given  as:  i  ~  (V%  —  V\}/rg. 
This  reading  of  grid  current 
includes  currents  from  the 
ionized  space  to  the  grid, 
leakage  currents,  and  grid 
emission.  If  the  test  is  made 
without  previous  operation 
of  the  tube,  the  grid-emission  FlG>  13^  cSxwak  Used  to  Check  Grid  Emission 

factor  will  be  eliminated. 

DEIONIZATION  TIME.     Deionization  time  of  a  gas  tube  is  the  time  required  for 
the  grid  to  gain  control  after  interruption  of  the  anode  current.  This  varies  with  condensed- 


4-68 


ELECTRON  TUBES 


mercury  temperature,  anode  current,  anode  voltage  immediately  after  discharge,  grid  volt- 
age, grid  circuit  impedance,  and  a  number  of  other  factors.    Not  only  is  the  peak  anode 

current  just  before  discharge  ceases 
important,  but  the  wave  shape  is 
also,  since  in  some  cases  the  ioniza- 
tion  due  to  an  earlier  peak  current 
may  decay  less  rapidly  than  the 
current,  thus  requiring  a  longer 
deionization  time.  Forcing  the 
anode  and  grid  negative  immedi- 
ately after  the  discharge  has  ceased 
decreases  the  deionization  time 


FIG.  14.     Circuit  Used  to  Check  Deionization  Time 


D-c 
supply 


FIG.  15.    Typical  Anode-cathode  Voltage  Trace  during 
Deionization  Time 


since  positive  ions  are  then  attracted  to  these  electrodes,  where  they  are  neutralized. 

To  restore  control  to  the  grid  it  is  not  necessary  to  remove  all  the  ions  from  the  grid- 
anode  space  but  only  to  reduce  the 
number  to  a  value  sufficiently  low  so 
that  the  grid  sheaths  overlap  enough  to 
prevent  discharge  when  a  positive  anode 
potential  is  again  applied.  Ions  may 
still  be  present  near  the  anode  or  near 
the  cathode,  with  the  grid  having  con- 
trol. This  shows  why  the  grid  voltage 
and  stiffness  of  the  grid  circuit  may  be 
relatively  more  important  in  deioniza- 
tion than  the  anode  circuit. 

Since    deionization    time    varies    so 
widely  with  all  these  variables,  it  is  es- 
sential to  know  all  these  conditions  under  which  a  measurement  has  been  made.    A  recom- 
mended circuit  for  making  such  tests  is  shown  in  Fig.  14.    The  operation  is  briefly  as  follows : 

The  tube  under  test  is  conduct- 
ing a  specified  d-c  current.  The 
capacitor  C  is  connected  in  parallel 
with  the  load  resistor  R  and  there- 
fore charges  so  that  the  negative 
capacitor  plate  is  connected  to  the 
tube  anode.  When  the  switch  S 
is  closed,  the  capacitor  makes  the 
anode  voltage  of  the  tube  instantly 
negative,  thus  stopping  the  dis- 
charge. The  capacitor  then  re- 
charges through  the  load  resistor 
until  the  anode  voltage  of  the  tube 
and  the  capacitor  voltage  are  equal 
to  the  supply  voltage.  The  rate  at 
which  this  anode  voltage  becomes 
positive  depends  upon  the  value  of 
the  capacitor  C  and  of  the  load 
resistor,  r.  These  values  can  easily 
be  used  to  calculate  the  length  of 
time  from  stopping  of  the  discharge 
until  the  anode  voltage  again  be- 
comes positive,  approximately  the 
point  at  which  the  tube  would 
conduct  if  deionization  is  not  com- 
plete. At  the  capacitor  setting  at 
which  the  tube  fails  to  control,  the 
deionization  time  can  be  calculated 
from  tfcp  formula  T  =  0.693rC. 
Any  desired  grid  conditions  may  be 
used  and  the  effect  of  these  factors 
determined.  Figure  15  shows  a 
typical  anode  voltage  trace  during 
the  test  operation. 


1000 
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Condensed  Hg  -  Temperature°C. 


80 


FIG. 


16.      Deionization   Time    Shown   as   a   Function  of 
Mercury  Temperature 


Other  modifications  of  this  circuit  may  be  used  to  allow  greater  variation  of  anode 
voltages,  or  inverter  circuits  may  be  set  up  which  will  operate  the  tubes  under  more  ex- 


PULSE  THYRATRONS 


4-69 


treme  conditions.    Figures  16,  17,  and  18  show  the  effect  of  some  of  the  factors  mentioned 
above. 


1200 
1100 
1000 
900 
800 

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»    500 
400 
800 
200 
100 
n 

Deionization  Time 
vs. 
Current  Density 
Grid  Bias       -150V. 
Grid  Resis.       500n 
Voltage             250V. 

Current  Density  thro  Baffle  Holes  -  A-mps./Sq,.  Irk 
FIG.  17.     Deionization  Time  Shown  as  a  Function  of  Tube  Current  Density 


1IUU 

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900 
800 

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Deionization  Time 
vs. 
Grid  Resistance 
Peak  Current       420  A 
Cond.  Angle            10° 
Voltage                 250V. 
Cond.  Hg  TemD.   30°C 

\ 

100     200     300    400     500    600     700     800    900    LQOO 

Grid  Resistance  in  Ohms 
FIG.  18.     Deionization  Time  Shown  as  a  Function  of  Grid  Eesistance 


18.  PULSE  THYRATRONS 

As  described  in  Section  9,  Pulse  Techniques,  many  circuits  have  been  developed  in 
which  the  currents  are  essentially  square-wave  pulses.  One  application  of  this  sort  pro- 
vides modulated  power  for  driving  a  magnetron;  here  a  thyratron  is  used  to  discharge  a 


4-70  ELECTRON  TUBES 

pulse-forming  line  or  network,  previously  charged  to  a  high  d-c  voltage,  through  a  pulse 
transformer  into  the  load.  The  thyratron  is  in  series  with  the  line  and  load,  and  it  carries 
a  pulse  current  of  magnitude  determined  by  the  voltage  charge  on  the  line  and  the  com- 
bined impedances  of  the  line  and  load. 

PULSE  VOLTAGES.  During  the  charging  of  the  line,  the  thyratron  is  subjected  to 
the  full  instantaneous  network  voltage,  so  that  it  must  be  capable  of  holding  off  this 
voltage.  It  is  generally  desirable  that  this  be  done  with  zero  grid  bias  on  the  tube,  and 
consequently  pulse  tubes  are  designed  so  that  they  will  fire  only  when  considerable  positive 
voltage  is  applied  to  the  grid.  This  is  generally  of  the  order  of  50  to  100  volts,  although  a 
higher  voltage  may  be  required  by  the  manufacturer's  data  in  order  to  insure  that  firing 
will  occur  with  a  minimum  of  time  variation. 

Time  jitter  is  a  common  term  for  expressing  the  variation  in  the  length  of  time  from 
application  of  the  grid  firing  voltage  until  start  of  the  anode  current  pulse.  Although  the 
actual  amount  of  this  delay  time  is  not  so  important,  any  variation  tends  to  destroy  the 
usefulness  of  systems  in  which  the  pulse  may  be  as  short  as  0.1  microsecond.  In  order  to 
keep  the  time  jitter  appreciably  lower  than  this  figure,  the  trigger  pulse  applied  to  the 
grid  should  have  a  steep  wave  front  and  a  peak  voltage  appreciably  higher  than  the  mini- 
mum required  to  cause  ionization.  The  internal  impedance  of  the  trigger  supply  must 
also  be  low  enough  to  provide  sufficient  grid  power  to  assist  in  accurate  firing. 

At  the  end  of  the  anode-current  pulse,  depending  on  the  ratio  of  load  and  line  imped- 
ances, the  voltage  across  the  line,  and  hence  across  the  tube,  may  become  negative.  The 
allowable  negative  voltage  is  limited  because  of  the  possibility  of  the  remaining  ionization 
causing  arcback.  After  deionization  is  complete,  the  inverse  voltage  may  be  increased, 
generally  to  about  the  forward  rating;  deionization  will  generally  occur  in  a  fairly  small 
percentage  of  the  total  time  of  application  of  inverse  voltage  and  will  allow  the  use  of  a-c 
charging  voltages,  other  factors  being  satisfactory. 

Tubes  are  tested  for  their  voltage  capacity  by  operation  in  a  pulse  circuit  at  the  required 
forward  voltage,  peak  pulse  current,  pulse  width  and  repetition  rate,  and  inverse  voltage, 
and  with  a  grid  circuit  which  represents  the  minimum  driving  conditions  which  may  be 
allowed. 

PULSE  CURRENTS.  Pulse  operation  for  thyratrons  was  not  originally  developed  for 
industrial  applications,  and  the  life  expectancies  under  such  operation  have  been  somewhat 
lower  than  those  of  other  uses  or  other  tubes.  Cathode  processing  requirements  are  dif- 
ferent, to  the  end  of  obtaining  from  10  to  20  times  the  peak  current  density  that  is  obtained 
from  industrial  tubes.  These  currents  are  limited  in  duration,  however,  to  a  pulse  width 
of  the  order  of  5  microseconds. 

Many  tubes  designed  for  industrial  operation  are  found  to  operate  satisfactorily  in 
low-power  pulse  circuits  at  much  higher  currents  than  their  nominal  ratings,  and  satisfac- 
tory pulse  ratings  have  been  developed  for  some  industrial  tubes. 

Pulse  tubes  are  generally  rated  for  a  maximum  pulse  repetition  rate,  pulse  width,  aver- 
age current,  and  duty.  For  service  in  which  none  of  these  factors  vary  with  time,  the 
pulse  width  (in  seconds)  times  the  repetition  rate  (in  pulses  per  second)  will  give  the  duty, 
or  fraction  of  the  total  time,  during  which  the  tube  is  actually  carrying  a  current  pulse, 
This  also  serves  as  a  relation  between  the  peak  and  average  currents,  and  thus  limits  the 
heating  of  the  tube  electrodes.  In  some  applications  it  may  be  desirable  to  operate  the 
tubes  at  a  varying  repetition  rate;  in  this  case  the  duty  is  sufficient  to  define  the  average 
current,  and  it  becomes  necessary  to  consider  the  minimum  amount  of  time  allowed  be- 
tween adjacent  pulses  to  take  care  of  deionization. 

Because  of  the  type  of  gas  used  in  order  to  meet  the  strict  deionization  requirements, 
and  the  current  densities  used,  the  tube  voltage  drop  during  the  pulse  is  inherently  high. 
In  addition,  ionization  requires  some  amount  of  time,  during  which  time  the  arc  voltage 
will  be  even  higher.  As  a  result,  the  tube  voltage  drop  may  rise  to  an  initial  value  of  100 
to  300  volts  and  then  drop  during  the  pulse  to  a  value  of  50  to  100  volts,  these  figures 
depending  on  the  type  of  tube  and  other  factors.  A  value  may  be  read  at  some  specific 
time  during  the  pulse  as  an  indication  of  tube  quality,  or  the  tube  voltage  may  be  plotted 
against  the  tube  current  on  an  oscilloscope,  the  shape  of  the  resultant  curve  serving  as  a 
basis  for  judgment.  In  this  case  the  voltage  is  generally  higher  during  the  rising  current 
than  during  the  falling  current,  and  the  area  enclosed  in  the  corresponding  traces  is  an 
indication  of  the  amount  of  ageing  of  the  cathode  during  conduction.  Information  for 
the  proper  testing  of  any  tube  type  may  be  requested  of  the  manufacturer. 
i*  CONSTRUCTION.  Although  some  industrial  types  of  tubes  have  found  pulse  appli- 
cations, these  have  been  at  low  repetition  rates  and  voltages,  and  most  of  the  usable  tubes 
have  been  designed  specifically  for  pulse  operation. 

The  cathodes  are  indirectly  heated,  since  filamentary  types  have  not  been  found  satisfac- 
tory. Heating  times  are  generally  in  the  order  of  1  to  3  min.  This  comparatively  short 


HOT-CATHODE  THYRATRON  TUBES  4-71 

heating  time  has  been  obtained  by  winding  light  heater  wire  either  inside  the  cathode 
cylinder  or  outside  a  cylinder  which  is  coated  on  the  inside  and  which  may  have  vanes 
projecting  toward  the  inside  to  increase  the  emitting  area.  The  alkali-earth  oxides  are 
used  for  the  emissive  coating. 

The  ionizing  medium  is  usually  hydrogen  because  of  its  light  weight  and  high  ion  mobil- 
ity and  consequent  speed  of  deionization.  Especial  care  must  be  taken  in  processing, 
however,  because  of  the  susceptibility  of  hydrogen  to  cleaning  up  into  the  parts,  particu- 
larly in  the  presence  of  contamination.  For  some  of  the  tubes,  high-purity  alloys  are  used 
which  are  not  required  in  the  ordinary  tube. 

The  anode  must  be  completely  shielded  from  any  possible  glow  discharge  or  arc  except 
the  normal  discharge,  and  it  is  surrounded  at  close  spacing  by  the  grid.  In  addition  to  the 
usual  openings  through  the  grid,  an  additional  shield  is  placed  directly  below  these  open- 
ings and  connected  to  the  grid  proper.  This  shield  assures  that  the  tube  will  maintain  its 
highly  positive  characteristic  and  operate  satisfactorily  at  high  voltage. 

Most  of  the  applications  have  been  met  by  three  tube  types,  one  operating  at  a  pulse 
voltage  of  3000  volts  and  a  pulse  current  of  35  amp;  the  second  at  8000  volts,  90  amp; 
and  the  third  at  16,000  volts  and  325  amp. 

19.  INSTALLATION  AND   OPERATION  OF  HOT-CATHODE  THYRA- 

TRON  TUBES 

For  any  particular  application  complete  data  should  be  obtained  from  the  tube  manufac- 
turer. Observation  of  the  instructions  will  be  amply  repaid  in  satisfactory  operation  and 
long  tube  life. 

FILAMENT  CIRCUIT.  The  greatest  single  cause  of  unsatisfactory  tube  operation  in 
the  past  has  been  incorrect  filament  voltage.  Too  low  a  filament  voltage  results  in  low 
emission  and  sputtering  of  the  cathode  which  will  give  very  short  tube  life  and  may  pre- 
vent the  tube  from  ever  operating  satisfactorily.  Too  high  a  filament  voltage  results  in 
an  excessive  rate  of  evaporation  of  cathode  material  which  results  in  short  tube  life.  A 
filament-voltage  variation  of  ±5  per  cent  is  generally  permitted.  Line-voltage  variations 
outside  the  specified  ±5  per  cent  limits  for  a  very  short  time,  such  as  5  or  10  sec,  are  not 
generally  very  harmful  in  the  cathode-type  tubes  where  the  cathode  temperature  changes 
rather  slowly.  If  excessive  voltage  variations  are  experienced  and  no  corrective  apparatus  is 
available,  it  is  sometimes  possible  to  put  the  apparatus  into  operation  by  so  adjusting  the 
filament  voltage  that  it  does  not  drop  more  than  5  per  cent  below  the  rated  value  during 
the  worst  periods  of  sustained  low  voltage.  Under  these  conditions  the  average  filament 
voltage  will  probably  be  high  and  tube  life  will  be  shortened  but  not  to  the  same  extent  as 
with  too  low  a  voltage. 

CATHODE  HEATING  TIME.  In  some  of  the  smaller  quick-heating  filamentary 
thyratrons  it  is  possible,  under  some  conditions  of  operation  when  the  anode  voltage  and 
current  are  rather  low,  to  apply  both  filament  and  plate  voltages  simultaneously  without 
causing  excessive  damage.  However,  when  full  rated  current  is  to  be  drawn  or  when  a 
slower  indirectly  heated  cathode  is  used,  it  is  essential  to  provide  some  means  whereby 
the  tube  will  not  pass  anode  current  until  the  cathode  is  up  to  operating  temperature. 
This  may  be  accomplished  by  a  switch  in  either  the  anode  or  load  circuit  or  by  proper 
bias  on  the  grid.  Wherever  it  can  be  economically  justified,  a  time-delay  relay  or  circuit 
may  be  used  to  give  this  protection.  Such  a  relay  will  also  provide  protection  in  the  event 
of  a  power  failure. 

TUBE  HEATING  TIME.  An  inert-gas-filled  thyratron  is  ready  to  operate  as  soon  as 
the  cathode  comes  up  to  temperature.  It  should  be  noted,  however,  that  in  a  mercury- 
vapor  tube  this  is  not  always  true.  The  condensed-mercury  temperature  must  be  within 
the  rated  limits  before  the  tube  is  operated.  Under  conditions  of  low  ambient  temperature 
where  it  is  desired  to  bring  the  tubes  up  to  operating  temperature  quickly,  tube  enclosures 
and  external  heaters  may  be  advantageous. 

INITIAL  HEATING.  During  shipment  of  mercury-vapor  tubes  the  mercury  may  have 
become  spattered  over  the  tube  electrodes.  When  first  put  into  service  the  tube  should  be 
allowed  to  heat  long  enough  to  evaporate  the  mercury  from  the  electrodes  and  to  distribute 
it  properly  before  anode  voltage  is  applied.  This  may  take  a  half  hour  or  longer,  depend- 
ing upon  the  size  of  the  tube. 

ANODE  CIRCUIT.  Tubes  should  not  be  used  in  circuits  where  the  voltages  are  higher 
than  the  rated  peak  inverse  or  peak  forward  voltages  of  the  tubes.  If  there  is  a  possibility 
that  transients  may  be  present,  a  cathode-ray  oscilloscope  or  calibrated  sphere  gap  across 
the  tube  may  prove  very  useful  in  determining  to  just  what  voltages  the  tube  is  sub- 
jected. 


4-72  ELECTRON   TUBES* 

CURRENT  OVERLOADS.  The  low,  essentially  constant,  voltage  drop  common  to 
gas-filled  tubes  makes  it  possible  to  overload  them  injuriously  to  a  greater  extent  than 
most  hard-vacuum  tubes,  which  have  a  rising  voltage  drop  with  increasing  current.  Once 
a  thyratron  tube  is  conducting,  the  current  is  limited  only  by  the  impedance  of  the  load 
circuit.  If  it  should  be  connected  to  a  very  low-reactance  source  of  power  with  no  load 
interposed,  an  abnormal  current  might  flow,  destroying  the  tube  in  a  fraction  of  a  second. 
A  number  of  tubes  are  ruined  in  this  way  by  experimenters,  particularly  those  who  are 
familiar  with  hard-vacuum  tubes  but  who  do  not  appreciate  fully  this  important  difference. 
For  this  reason  it  is  very  important  to  make  sure  that  some  current-limiting  impedance  is  in 
series  with  both  the  anode  and  grid  leads  when  voltage  is  applied. 

This  low,  constant  voltage  drop  makes  it  possible  to  impose  heavy  peak  current  over- 
loads on  the  tubes  without  drawing  excessive  average  current  as  indicated  by  a  d-c  ammeter 
directly  in  series  with  the  tube.  Unless  sufficient  reactance  is  present  in  the  circuit,  these 
peaks  may  be  experienced  when  tubes  are  used  in  rectifier  applications  with  capacitors 
connected  directly  across  the  output,  when  tubes  are  used  to  charge  storage  batteries,  or 
when  they  are  operating  into  any  type  of  counter  emf  load.  Although  calculations  may 
sometimes  be  made,  it  is  generally  safest  to  check  tube  currents  with  an  oscilloscope  when 
high  peak  currents  are  suspected. 

REACTANCE  OF  POWER  SOURCE.  It  is  desirable  to  include  enough  reactance  in 
the  anode  transformers  or  other  sources  of  power  to  limit  the  tube  short-circuit  currents 
under  any  type  of  fault  or  failure  to  a  value  not  greater  than  the  surge  current  rating  of 
the  tube  and  to  provide  protective  fuses  or  other  interrupting  devices  to  open  the  circuit  in 
a  reasonably  short  time,  generally  not  in  excess  of  0.1  sec.  Experience  has  shown  that 
tubes  operate  much  more  satisfactorily  in  circuits  in  which  this  precaution  is  taken  than 
in  ones  in  which  no  such  protection  is  provided.  Various  theories  have  been  offered  in 
explanation,  but  the  fact  has  been  well  proved  that  high-reactance  circuits  give  long  tube 
life  and  low-reactance  circuits  give  short  tube  life.  In  multitube  circuits  arcback  of  one 
tube  often  overloads  the  others,  and  again  the  circuit  reactance  must  be  depended  upon 
to  prevent  permanent  injury  to  these  tubes.  If  excessively  low-reactance  transformers 
are  used  and  if  tube  failures  are  encountered  which  show  evidence  of  heavy  overloads,  such 
as  blown-up  stems  or  cracked  seals,  a  small  resistance  or  inductance  placed  directly  in  the 
anode  lead  of  each  tube  will  often  stop  the  trouble. 

GRID  CIRCUIT.  Usually  the  source  of  grid  power  itself  has  sufficient  impedance  to 
prevent  excessive  grid  current  from  damaging  the  tube.  If  any  doubt  is  felt  on  this  score 
a  grid  resistor  should  be  used  to  prevent  accidents.  However,  in  certain  types  of  inverter 
circuits  it  is  advisable  to  use  a  relatively  stiff  grid  circuit  to  help  speed  up  deionization. 
In  the  selection  of  a  resistor  the  grid-current  specification  of  the  particular  tube  to  be  used 
should  be  considered  so  that  difficulty  will  not  be  experienced  through  the  loss  of  bias 
caused  by  the  flow  of  grid  current  in  the  grid  resistor.  This  resistor  should,  however,  be 
large  enough  so  that  the  voltage  measured  directly  at  the  grid  with  the  tube  conducting 
does  not  exceed  its  rated  value  with  the  normal  operating  grid-supply  voltage. 

20.  COLD-CATHODE  TUBES 

The  term  "cold-cathode  tube"  refers  to  tubes  in  which,  in  general,  the  discharge  is  the 
self-sustaining  glow  type  previously  described. 

Since  110  cathode  heating  is  required,  the  cathode  life  is  not  affected  by  standby  opera- 
tion, and  these  tubes  are  particularly  adapted  to  relay  circuits  of  infrequent  operation,  if 
the  current  requirement  is  sufficiently  small,  in  addition  to  the  familiar  regulator  and 
rectifier  applications. 

Cold  Cathode.  A  cold  cathode  is  a  cathode  operating  at  a  temperature  at  which 
thermionic  emission  is  negligible. 

Starter  (or  control  anode).  A  starter  of  a  cold-cathode  tube  is  an  electrode  ordinarily 
used  in  conjunction  with  a  cathode  to  initiate  conduction 

Control  Gap.  A  control  gap  of  a  cold-cathode  tube  is  the  conduction  path  between  a 
starter  and  cathode  in  which  conduction  is  ordinarily  initiated. 

Main  Gap.  A  main  gap  of  a  cold-cathode  tube  is  the  conduction  path  between  a 
cathode  and  a  principal  anode  in  which  the  principal  conduction  ordinarily  takes 
place. 

Anode  Breakdown  Voltage.  The  anode  breakdown  voltage  of  a  cold-cathode  tube  is 
the  anode  voltage  required  to  cause  conduction  to  take  place  in  the  main  gap  when  the 
control  gap  is  not  conducting. 

Starter  Breakdown  Voltage.  The  starter  breakdown  voltage  of  a  cold-cathode  tube  is 
the  starter  voltage  required  to  cause  conduction  to  take  place  in  the  control  gap. 


COLD-CATHODE  TUBES 


4-73 


Transfer  Current.  The  transfer  current  of  a  cold-cathode  tube  is  the  control-gap  cur- 
rent required  to  cause  conduction  in  the  mam  gap  with  positive  voltage  applied  to  the 
anode. 

Regulation.  Regulation  of  a  cold-cathode  tube  is  the  difference  in  voltage  drop  ob- 
tained over  a  range  of  conducted  current. 

Inverse  Anode  Current.  The  inverse  anode  (or  starter)  current  is  the  electron  current 
flowing  from  the  associated  circuit  to  the  anode  (or  starter)  of  a  cold-cathode  tube. 

Anode  Drop.  The  anode  drop  of  a  cold-cathode  tube  is  the  main-gap  voltage  drop 
after  conduction  is  established  in  this  gap. 

Starter  Drop.  The  starter  drop  of  a  cold-cathode  tube  is  the  control-gap  voltage  drop 
after  conduction  is  established  in  this  gap. 

VOLTAGE  REGULATOR  TUBES.  Two-electrode,  inert-gas-filled  cold-cathode  tubes 
are  generally  operated  within  a  certain  range  of  d-c  load  currents  within  which  the  tube 
voltage  drop  remains  essentially  constant.  For  instance,  such  a  tube  might  break  down 
or  become  ionized  with  a  potential  of  125  volts  on  the  anode  and  then  operate  within  3  or 
4  volts  of  a  90-volt  tube  drop  within  the  specified  current  range.  As  with  the  hot-cathode 
tubes,  the  variation  in  supply  voltage  therefore  appears  across  the  load  resistance,  and  it  is 
essential  that  this  resistance  be  sufficient  to  maintain  the  load  current  within  the  required 
value.  Some  calculations  are  generally  necessary  to  insure  that  adequate  starting  voltage 
is  also  available  when  the  supply  voltage  to  be  regulated  falls  to  its  minimum,  value. 

Since  regulator  tubes  are  designed  to  conduct  current  in  one  direction  only,  they  gen- 
erally have  one  small  electrode  and  one  large  electrode,  the  large  one  acting  as  the  cathode. 
Tube  life  depends  upon  the  current  density  on  the  cathode  (and  therefore  the  size  of  the 
cathode),  since  sputtering  of  the  metal  will  occur  if  the  element  is  overloaded. 

THREE-ELECTRODE  COLD-CATHODE  TUBES.     A  third  electrode  or  starter  may 
be  introduced  into  a  cold-cathode  tube  to  control  the  starting  of  the  discharge.    As  with 
hot-cathode  tubes  the  starter  has  no  appreciable  control  once  the  discharge  has  occurred 
in  the  main  gap  until  the  anode  potential  has 
dropped  below  the  tube  voltage  drop  (some- 
times called  sustaining  voltage). 

The  use  of  these  tubes  as  relays  is  obvious 
because  of  their  control  characteristics.  An- 
other property  of  such  a  tube  is  that,  although 
some  inverse  anode  current  will  be  conducted 
with  negative  anode  voltage,  the  breakdown 
voltage  in  this  direction  is  somewhat  high 
and  the  anode  drop  increases  rapidly  as 
current  increases.  Figure  19  shows  this 
characteristic.  With  the  starter  connected 
to  the  anode  through  a  high  resistance,  the 
forward  breakdown  voltage  will  be  much 
below  the  normal  main-gap  breakdown  value 
and  the  tube  will  operate  satisfactorily  as  a 
rectifier.  The  circuit  must  be  such,  of  course, 
that  peak  inverse  anode  current  or  voltage 
ratings  are  not  exceeded. 

The  starter  drop  is  quite  independent  of 
control-gap  current,  and  the  tubes  are  used 
as  voltage  regulators  with  the  cathode  and 
starter  as  electrodes. 


80 


40 


- 


eo 


-160 


-200 


-30    -20   -10       0        10       20      30      40 
Anode  current,  mlUiamperes 


FIG.  19. 


Typical  Cold-cathode-tube  Character- 
istic 


The  mechanism  of  operation  of  the  starter  is  similar  to  that  of  the  grid  in  a  positive-grid 
thyratron.  When  positive  voltage  is  applied  to  the  starter,  a  small  electron  current  known 
as  a  Townsend  current  will  flow  from  the  cathode.  As  the  voltage  and  the  Townsend  cur- 
rent are  increased,  a  point  is  reached  at  which  ionization  occurs  in  the  control  gap.  The 
control-gap  current  may  then  be  increased  by  the  control  circuit  until  the  transfer  current 
is  reached,  at  which  point  (depending  upon  the  anode  voltage)  conduction  occurs  to  the 
anode.  At  the  anode  breakdown  voltage,  conduction  will  occur  without  ionization  in  the 
control  gap,  and  therefore  the  transfer  current  is  zero.  At  the  other  extreme,  the  tube  will 
not  sustain  a  discharge  at  less  than  the  anode  drop,  and  hence  the  transfer  current  required 
at  this  point  is  infinite.  Figure  20  shows  a  typical  transfer-current  characteristic.  It  is 
apparent  that  the  anode  operating  voltage,  when  starter  control  is  expected,  must  lie 
somewhere  between  the  anode  drop  and  the  anode  breakdown  voltage. 

The  control  characteristics  are  subject  to  variation  during  life  but  may  be  expected  to 
remain  constant  within  5  or  6  volts  during  most  of  life.  Variation  during  shelf  life  may 
*be  of  the  same  order  of  magnitude  depending  upon  the  length  of  storage  and  the  light 


4-74 


ELECTRON  TUBES 


conditions  for  those  tubes  which  do  not  have  an  opaque  coating.  The  effect  of  light  is 
negligible  for  medium  light  levels  but  will  be  appreciable  at  levels  approaching  darkness 
or  direct  sunlight,  the  numerical  value  of  breakdown  voltages  decreasing  as  light  intensity 
is  increased,  and  conversely  with  decreased  intensity.  Most  variations  occurring  during 
storage  will  disappear  after  a  few  seconds  of  operation. 


170 
160 

150 

140 
S 
£130 

JS120 

g 

I  110 

100 
90 
80 
70 


8        10      12      14       16      18 
Transier  current,  microamperes 


20      22        24 


FIG.  20.     Typical  Transfer  Current  Characteristic 

TEST  FOR  COLD-CATHODE  TUBES.  Because  of  the  effect  of  light  and  storage 
upon  control  characteristics,  it  is  generally  advisable  that  all  tests  be  made  with  moderate 
illumination  and  with  a  few  seconds  of  current  conduction  immediately  before  the  test  in 
order  to  stabilize  the  readings.  Because  of  the  low  values  of  transfer  current  some  errors 
may  be  introduced  if  capacitance  effects  are  ignored,  and  it  is  therefore  important  that 
the  starter  resistance  be  placed  immediately  adjacent  to  the  starter  electrode. 

Breakdown  voltage  is  measured  by  applying  a  positive  d-c  voltage  to  the  anode  and 
increasing  it  until  the  tube  conducts  current.  The  minimum  value  of  voltage  required  to- 
start  conduction  is  measured.  After  the  tube  conducts,  the  tube  drop  may  be  measured  at 
specified  current  values  and  the  regulation  thereby  determined.  For  three-electrode  tubes 
the  anode  breakdown  voltage  is  measured  as  above,  except  with  the  starter  connected  to- 
the  anode  through  a  resistance  generally  not  exceeding  50,000  ohms. 

Transfer  current  is  measured  by  applying  the  specified  d-c  anode  potential  and  a  positive 
starter  voltage,  with  sufficient  starter  resistance  to  limit  the  starter  current  to  less  than 
the  transfer  value.  The  starter  current  is  then  increased  until  conduction  takes  place  in 
the  main  gap,  at  which  point  the  transfer  current  is  measured  by  means  of  a  microammeter 
in  series  with  the  starter.  Care  must  be  exercised  to  insure  that  the  starter  is  electrically- 
connected  at  every  instant  during  this  test. 

STROBOTRON  TUBES.  A  modification  of  the  three-electrode  cold-cathode  tube 
described  is  commercially  called  the  Strobotron.  This  tube  commonly  has  two  control 
electrodes  or  grids  together  with  the  cathode  and  anode.  The  cathode  is  coated  with  a, 
cesium  compound,  which  breaks  down  during  conduction  and  liberates  free  cesium.  The- 
tube  is  inert-gas-filled,  and  conduction  occurs  by  formation  of  a  cathode  spot  on  the  metallic 
cathode  surface  by  concentration  of  a  glow  discharge.  The  voltage  drop  is  therefore  much 
lower  than  that  of  a  glow  discharge.  These  tubes  are  commonly  used  with  neon  gas  as 
a  source  of  high-intensity  light  for  stroboscopic  purposes  but  may  also  be  used  for  control 
or  relay  purposes.  Capacity  is  limited  by  cathode  heating  during  conduction  and  by  ion 
bombardment  of  the  cathode  as  with  other  cold-cathode  tubes.  For  this  reason,  although 
very  high  peak  currents  can  be  handled,  the  average  current  rating  is  generally 
quite  low,  £$£ 

A  capacitance  is  generally  used  across  the  tube  in  operation.  This  capacitance  is  dis- 
charged through  the  tube  in  order  to  provide  very  high  instantaneous  current.  It  also 
serves  as  a  means  of  stopping  the  discharge,  since,  as  the  capacitance  discharges  to  the 
tube  drop,  the  conduction  changes  to  a  glow  rather  than  an  arc.  This  immediately  raises 
the  tube  drop  and  the  conduction  ceases,  the  tube  becoming  deionized  before  the  capacitor 
recharges  to  sufficient  voltage  to  maintain  even  a  glow  discharge.  Operation  at  too  high. 


POOL-CATHODE  TUBES 


4-75 


a  frequency,  or  -with  improper  circuit  conditions,  will  frequently  maintain  a  glow  discharge 
in  the  tube  and  lose  the  advantage  of  the  tube's  capacity  for  high  peak  currents. 

With  this  tube  as  with  other  cold-cathode  tubes  life  is  a  function  of  average  current  and 
conduction  time,  there  being  no  appreciable  deterioration  during  storage. 

Available  types  of  tubes  are  shown  in  Table  2. 

Table  2.    Available  Types  of  Cold-cathode  Tubes 


Type 

Desig- 
nation 

Cathode  Current, 
milliamperes 

Anode 
Voltage 
Drop 

Breakdown 
Voltage 
s  —  starter 
a  =  anode 

Remarks 

Average 

Peak 

874 

RCA 

10-50 

90 

125a 

Voltage  regulator 

991 

RCA 

2 

3 

4&-67 

87a 

Voltage  regulator 

BR 

Ray 

50 

60 

Rectifier 

OA2 

S,  Hy, 

5-30 

150 

I55a 

Miniature  voltage  regulator 

RCA 

OB  2 

Hy 

5-30 

105 

133a 

Miniature  voltage  regulator 

OA3/VR75 

GL-,  S, 

5-40 

75 

lOOa 

Voltage  regulator 

RCA 

OB3/VR90 

GL-,  S 

10-30 

90 

HOa 

Voltage  regulator 

OC3/VR105 

GL-,  S,  Hy, 

5-40 

105 

11  5o 

Voltage  regulator 

RCA 

OD3/VR150 

GL-,  S, 

5-40 

150 

I60a 

Voltage  regulator 

RCA,Hy 

1B46 

S 

1-  2 

79-85 

225a 

Miniature  voltage  regulator 

1B47 

S 

1-  2 

75-90 

225a 

Miniature  voltage  regulator 

1B64 

S 

1-  2 

65-75 

225a 

Miniature  voltage  regulator 

31  3C 

WE 

35 

100 

75  at  20  ma 

70s 

Control  tube 

313CC 

WE 

18 

50 

75  at  20  ma 

72s 

Control  tube 

359A 

WE 

15 

40 

75  at  10  ma 

75s 

Control  tube 

395A 

WE 

13 

35 

75  at  10  ma 

77s 

Control  tube 

OA4G 

S,  RCA 

25 

100 

70 

75-90s 

Control  tube 

1C21 

RCA 

25 

100 

73 

66-80s 

Control  tube 

618 

KU- 

15 

100 

180 

Control  tube 

1B48 

Ray 

6 

50 

1  00  at  6  ma 

800a 

High-voltage  rectifier 

Prefix  Used  by 

GL-        General  Electric  Company 

Westinghouse  Electric  Corporation 


KU- 

Letter 

WE 

RCA 

Ray 

S 

Hy 


Indicates 

Western  Electric  Company 
Radio  Corporation  of  America 
Raytheon 

Sylvania  Electric  Products,  Inc. 
Hytron  Radio  and  Electronics  Corporation 


21.  POOL-CATHODE  TUBES 

Pool  Cathode.  A  pool  cathode  is  a  cathode  in  which  the  principal  source  of  electron 
emission  is  a  cathode  spot  on  a  metallic  pool  electrode. 

Cathode  Spot.  A  cathode  spot  is  an  area  on  the  cathode  of  an  arc  from  which  electron 
emission  takes  place  at  a  current  density  of  thousands  of  amperes  per  square  centimeter 
and  where  the  temperature  of  the  electrode  is  too  low  to  account  for  such  currents  by 
thermionic  emission. 

Pool  Tube.     A  pool  tube  is  a  gas  tube  with  a  pool  cathode. 

Single-anode  Tube.     A  single-anode  tube  is  an  electron  tube  having  a  single  main  anode. 

Multianode  Tube.  A  multianode  tube  is  an  electron  tube  having  two  or  more  main 
anodes  and  a  single  cathode. 

Ignitron.  An  ignitron  is  a  single-anode  pool  tube  in  which  an  ignitor  is  employed  to 
initiate  the  cathode  spot  for  each  conducting  period. 

Excitron.  An  excitron  is  a  single-anode  pool  tube  provided  with  means  for  maintaining 
a  continuous  cathode  spot. 

Pumped  Rectifier.  A  pumped  rectifier  is  a  rectifier  which  is  continuously  connected 
to  evacuating  equipment  during  operation. 

Sealed  Tube.     A  sealed  tube  is  a  tube  which  is  hermetically  sealed  after  degassing. 

Ignitor.  An  ignitor  is  a  stationary  electrode  which  is  partially  immersed  in  the  cathode 
pool  and  has  the  function  of  initiating  a  cathode  spot. 


4-76 


ELECTRON  TUBES 


As  mentioned  previously,  tubes  of  the  pool-cathode  type  make  use  of  the  self-sustaining 
form  of  discharge.  The  cathode  dark  space  is  very  small  because  of  the  high  current  den- 
sity, and  the  resulting  high  voltage  gradient  is  assumed  to  cause  field  emission  and  thus 
maintain  the  high  current  density  at  an  arc  voltage  drop  near  that  of  the  ionization  voltage. 

Since  there  is  no  emissive  material  to  be  damaged,  the  pool  cathode  is  capable  of  carrying 
extremely  high  currents  without  any  deterioration.  Tubes  havdng  such  cathodes  can 
therefore  be  used  in  applications  where  very  high  peak  currents  are  required  for  short 
periods  of  time,  or  where  it  is  desirable  that  a  high  short-circuit  or  arcback  current  be 
allowed  without  damage  to  the  tube.  This  current  can  be  much  higher  proportionally 
than  for  a  hot-cathode  tube.  The  pool  cathode  also  has  the  advantage  of  not  requiring 
heating  time. 

On  the  other  hand,  the  pool  tube  requires  auxiliary  circuits  for  maintaining  ionization 
or  for  starting  the  ionization  during  each  operating  cycle.  The  size  of  the  tube  and  the 
corresponding  volume  of  ionization  increases  the  difficulty  of  building  high-voltage  tubes, 
or  those  with  rapid  deionization.  Further,  because  of  the  amount  of  current  capacity, 
water  cooling  is  required  for  most  types,  and  corresponding  protection  must  be  provided 
to  insure  proper  water  flow  and  water  temperature  at  all  times. 

Available  types  of  pool  tubes  are  listed  in  Table  3. 

Table  3.    Available  Pool-cathode  Tubes 


Type 


Designation 


Typical  Ratings 


415 

681/686 

652/657 

271 

235-A 

651/656 

655/658 

258-A 

259-B 

679 

653-B 

238-B 

ES-8-01 

507 

688 

427 
506 

8-in.  tank 
10-in.  tank 

1 0-in.  tank 
1 2-in.  tank 

6-EP-20-01 
1 6-in.  tank 

16-in.  tank 
15-in.  tank 

20-in.  tank 
2  2-in.  tank 
6-EP-20-11 
HF-26 


GL- 
WL- 
WL- 
GL- 
GL- 
WL- 
WL- 
GL- 
GL- 

WL- 

WL- 

GL- 

Allis-Chalmers 

GL- 

WL- 

GL- 
GL- 

Westinghouse 
General  Electric 

General  Electric 
Westinghouse 

Allis-Chalmers 
General  Electric 

General  Electric 
Westinghouse 

General  Electric 
Westinghouse 
Allis-Chalmers 
Allis-Chalmers 


300-kva,  22.4-amp  average,  welder  control  ignitron  (sealed) 

300-kva,  22.4-amp  average,  welder  control  ignitron  (sealed) 

600-kva,  56-amp  average,  welder  control  ignitron  (sealed) 

600-kva,  56-amp  average,  welder  control  ignitron  (sealed) 

1200-kva,  140-amp  average,  welder  control  ignitron  (sealed) 

1 200-kva,  140-amp  average,  welder  control  ignitron  (sealed) 

2400-kva,  355-amp  average,  welder  control  ignitron  (sealed) 

2400-kva,  355-amp  average,  welder  control  ignitron  (sealed) 

600  volts  d-c,  200  kw  (output  of  6  tubes),*  rectifier  ignitron  (sealed) 

250  volts  d-c,  150kw 

600  volts  d-c,  200  kw  (output  of  6  tubes),*  rectifier  ignitron  (sealed) 

250  volts  d-c,  150kw 

600  volts  d-c,  500  kw  (output  of  6  tubes),*  rectifier  ignitron  (sealed) 

250  volts  d-c  300  kw 

600  volts  d-c,  500  kw  (output  of  6  tubes),*  rectifier  ignitron  (sealed) 

250  volts  d-c,  300  kw 

600  volts  d-c,  500  kw  (output  of  6  tubes),*  rectifier  excitron  (sealed) 

250  volts  d-c,  300  kw 

600  volts  d-c,  1000  kw  (output  of  6  tubes),*  rectifier  ignitron  (sealed) 

250  volts  d-c,  600  kw 

600  volts  d-c,  1000  kw  (output  of  6  tubes),*  rectifier  ignitron  (sealed) 

250  volts  d-c,  600  ]|cw 

350-volt  peak  1 0-amp  glass  demonstration  ignitron  (sealed) 

9000-volts  d-c,   8100  kw  (output  of  6  tubes),*  rectifier  and  inverter 

ignitron  (sealed) 

600-volts  d-c,  1000  kw  (output  of  6  tanks),*  rectifier  ignitron  (pumped) 
600-volts  d-c,  1000  kw  (output  of  6  tanks),*  rectifier  ignitron  (pumped) 
250-volts  d-c,  625  kw 

3000  volts  d-c,  1 500  kw  (output  of  6  tanks),*  rectifier  ignitron  (pumped) 
600  volts  d-c,  1500  kw  (output  of  6  tanks),*  rectifier  ignitron  (pumped) 
250  volts  d-c,  750  kw 

600  volts  d-c,  1500  kw  (output  of  6  tanks),*  rectifier  excitron  (pumped) 
250  volts  d-c,  750  kw 

600  volts  d-c,  2000  kw  (output  of  6  tanks),*  rectifier  ignitron  (pumped) 
250  volts  d-c,  lOOOkw 

3000  volts  d-c,  4000  kw  (output  of  6  tanks),*  rectifier  ignitron  (pumped) 
600  volts  d-c,  2000  kw  (output  of  6  tanks),*  rectifier  ignitron  (pumped) 
250  volts  d-c,  lOOOkw 

600  volts  d-c,  3000  kw  (output  of  6  tanks),*  rectifier  ignitron  (pumped) 
250  volts  d-c,  1 500  kw 

600  volts  d-c,  3000  kw  (output  of  6  tanks),*  rectifier  ignitron  (pumped) 
250  volts  d-c,  1 500  kw 
1750  volts  d-c,   2500  kw  (output  of  6  tanks),*  rectifier  and  inverter 

excitron  (pumped) 
1000  cycles,  300  kw,  multianode  frequency  changer  (pumped) 


*  These  are  typical  operating  conditions  rather  than  absolute  maximum  ratings. 
WL-,  prefix  used  by  Westinghouse  Electric  Corporation;  GL-,  prefix  used  by  General  Electric  Com- 
pany. 


POOL-CATHODE  TUBES 


4-77 


CLASSIFICATION.  Pool-cathode  tubes  may  be  classified  as  sealed  or  pumped  units. 
In  general,  the  single-anode  sealed  units  are  of  comparatively  lower  current  capacity,  now 
being  built  in  the  range  from  about  50  amp  average  to  about  400  amp  average.  After  the 
initial  exhaust  treatment  these  tubes  are  permanently  sealed  and  no  further  pumping  is 
required.  Pumped  units  have  been  built  largely  of  metal  in  either  single-anode  units  or 
in  a  multianode  form  commonly  called  a  tank.  These  units  have  a  pump  or  system  of 
pumps  operating  to  maintain  vacuum.  Current  capacity  of  these  tanks  is  in  the  order  of 
1000  amp  average.  Pool  tubes  may  be  classified  still  further  as  tubes  in  which  there  is 
continuous  ionization  and  those  in  which  the  arc  is  initiated  each  time  conduction  starts 
and  is  extinguished  at  the  end  of  each  conducting  period. 

MERCURY-ARC  RECTIFIERS.  The  mercury-arc  rectifier  is  a  familiar  example  of 
a  pool  tube  in  which  continuous  ionization  is  maintained. 


FIG.  21.     Typical  Pumped  Mercury-arc  Rectifier  Tank.     (Courtesy  of  G.  E.  Co.) 


These  types  are  made  in  both  glass  and  metal.  The  glass  types  are  sealed  units  with 
the  mercury  pool  in  the  bottom  and  with  glass  arms  extending  out  from  the  main  body  of 
the  tube,  slightly  above  the  pool  level.  An  anode  is  sealed  into  the  end  of  each  arm,  which 
may  or  may  not  be  bent,  depending  on  the  inverse  voltage  rating  of  the  tube.  Some  two- 
anode  types  are  made  for  lighting  and  railway  service,  but  the  more  extensive  applica- 


4-78 


ELECTRON  TUBES 


Ferntco  Metal  Alloy 

and  Pyrex  Type 

Glass  Seal 


tions  are  in  three-  or  six-anode  designs.  These  applications  are  mainly  outside  the  United 
States  and  are  in  railway  service  and  other  power  usage.  A  typical  six-anode  tube  wou]ld 
have  a  capacity  of  the  order  of  400  amp  average  at  600  volts. 

The  arc  spot  of  such  tubes  is  generally  formed  by  tilting  the  tube  so  that  the  mercury 
pool  forms  a  circuit  with  a  starting  electrode.  When  this  circuit  is  broken  an  arc  is  ini- 
tiated and  is  maintained  by  one  or  more  auxiliary  anodes.  These  anodes  form  with  the 
pool  a  low-voltage  rectifier  the  purpose  of  which  is  to  keep  enough  current  (6  or  7  amp) 
flowing  through  the  tube  to  maintain  the  cathode  spot.  As  the  tube  current  increases, 
the  cathode  spot  breaks  up  into  several  spots  each  carrying  about  the  same  amount  of 
current. 

These  tubes  may  be  given  grid-control  characteristics  by  the  addition  of  a  grid  around 
the  anode  in  each  arm. 

Figure  21  shows  an  example  of  a  metal  multianode  pumped  rectifier  tank  rated  for 
1000  kw  at  600  volts.  Such  tanks  operate  like  the  glass  tubes  except  that  the  arc  is  ini- 
tiated by  a  probe  electrode  in  the  center  of  the  pool.  This  electrode  is  magnetically  con- 
trolled and  makes  and  breaks  contact  with  the  pool  in  order  to  form  the  arc  spot.  loniza- 
tion  is  then  maintained  by  auxiliary  anodes  such  as  the  one  shown  on  the  right-hand  side 
of  the  tank,  extending  from  the  top  cover  into  the  main  volume.  One  of  the  main  anodes 

is  shown  at  the  left,  with  a 
suitable  grid  structure  sur- 
rounding and  with  addi- 
tional baffling  below.  The 
bottom  surface  of  the  tank 
slopes  to  allow  the  return  of 
condensed  mercury  to  the 
pool  in  the  center,  and  a 
water-cooling  system  ex- 
tends completely  around  the 
outside  of  the  tank,  under 
the  mercury  pool,  and 
through  a  cooling  coil  and 
cylinder  in  the  center  of  the 
tank.  This  provides  a  maxi- 
mum of  cooling  area  in  con- 
tact with  the  arc  and  thus 
provides  the  best  control 
of  operation.  Direction  of 
water  flow  is  indicated  by 
arrows  in  the  cooling  sys- 
tem. The  exhaust  connec- 
tion, vacuum  controls,  and 
pumps  are  shown  at  the  top 
and  right  of  the  tank. 

Such  tanks  are  built  with 
voltage  ratings  as  high  as 
3000  volts  direct  current 
and  with  currents  of  the 
order  of  1000  amp. 

IGNITRONS.  Figure  22 
shows  a  typical  sealed  igni- 
tron  tube.  The  two  enclos- 
ing cylinders  provide  a  path 
for  cooling  water,  and,  since 
the  temperature  of  the  inner 
tube  wall  is  important  in 


Flow-dlreciiO] 
Vanes 


Delonization  Baffle 


Splash-hood  Baffli 


Tube  Support  and 
'Cathode  Connection 


Main  Graphite  Anode 


Starting  Igniters 


•Mercury  PooJ. 
Cathode 


'Seal-off7* 


PIG.  22.     Sealed  Ignitron  for  Power  Rectifier  Service, 
of  G.  E.  Co.) 


(Courtesy 


deternuning  mercury-vapor  pressure  and  operating  capacity  of  the  tube,  it  is  essential 
that  the  required  water  flow  be  provided  during  operation.  The  graphite  anode  is 
designed  for  a  maximum  transfer  of  heat  from  its  face  and  for  a  minimum  of  heat  genera- 
tion from  the  high  current  flow.  Mercury  is  thrown  up  from  the  pool  by  the  action  of 
the  cathode  spots,  and  a  splash-hood  baffle  is  placed  directly  above  the  center  of  the 
cathode  to  prevent  splashes  of  mercury  from  hitting  the  anode  or  upper  tube  walls  and 
thereby  increasing  the  possibility  of  arcback.  The  purpose  of  a  deionization  baffle  is 
easily  seen,  since  it  reduces  the  distance  from  the  ionized  ai;ea  to  a  metal  part.  This  part 
is  particularly  important  in  tubes  designed  for  rectifier  service,  where  the  deionization 
requirements  are  more  severe  than  in  welder  use.  The  auxiliary  anode,  also,  is  used  for 


POOL-CATHODE  TUBES 


4-79 


rectifier  service,  where  it  is  desirable  to  maintain  ionization  over  the  conducting  period  in 
order  to  provide  for  main  anode  currents  below  the  value  at  which  the  arc  becomes  un- 


1  MYCALEX     ANODE    INSULATOR 

2  MYCALEX  INSULATOR  (LEAD  TO  INSULATED  BAFFLE) 

3  ANODE   HEATER    COVER 

4  ANODE   HEATER 

5  VACUUM  CHAMBER • COVER 

€  TWO  (2)   ALUMINUM    GASKETS 

7  VACUUM    CHAMBER 

8  WATER   JACKET 

9  GRID 

10  SUPPORT   RING    FOR    PT.  9 

I  I    INSULATOR    FOR    PT   9    (  MYCALEX) 

I  2  GRAPHITE    ANODE 

I  3  ANODE     STUD 

I  4  MERCURY   SPLASH    BAFFLE 

I  5  IGNITOR    TIP 

I  6  INDIVIDUAL  VACUUM    VALVE 

17  HEAT    SHIELD 

18  ANODE  SPACER 

I  9  MYCALEX   INSULATOR  FOR  IGNITOR   8  REUEVMG  ANODE  LEADS 

20  ADJUSTING  SCREWS   FOR   PT.  15 

2 1  FLEXIBLE   DIA   {AND  ADJUSTING  SCREWS   PT  2( 

PERMIT   ADJUSTMENT  Of  1GNITOR  IMMEF " 

POOL)  TO  CORRECT  VALUE    FROM    OUTSI 

22  RELIEVING    ANODE  FOR   JGNITOR    TIP 

23  MERCURY   SEPARATORS 

24  MERCURY    POOL   (CATHODE) 

25  CATHODE    CONNECTION 
ZS  EXHAUST   BAFFLE 

If  EXHAUST    PIPE 
28  GASKETS  (INNER  -  FORM VAR) 
,C  OUTER -ALUMINUM) 


FIG.  23.    Pumped  Ignitron  Tank.     (Courtesy  of  G.  E.  Co.) 

stable,  usually  3  to  4  amp.  Tubes  for  welder  applications  may  be  built  without  the  two 
baffles  and  the  auxiliary  anode.  On  the  other  hand,  tubes  built  for  high-voltage  rectifier 
use  may  have  several  baffles  or  grids  surrounding  the  anode  in  order  to  provide  high- voltage 


4-80  ELECTRON  TUBES 

control  and  to  reduce  deionization  time,  and  they  may  also  have  two  or  more  ignitors  in 
order  to  assure  operation  in  the  event  of  an  ignitor  failure. 

Figure  23  shows  a  typical  pumped  ignitron  tank,  with  vacuum-tight  joints  and  flexible 
ignitor  support  to  allow  adjustment  of  the  ignitor  position;  the  other  parts  are  similar  in 
use  to  those  in  the  sealed  tube.  The  sealed  and  pumped  units  are  used  in  similar  applica- 
tions. 

IGNITOR  CHARACTERISTICS.  The  ignitor  as  shown  in  Fig.  22  is  shaped  somewhat 
in  the  form  of  a  pencil  and  is  composed  largely  of  boron  carbide.  The  composition  is  such 
that  the  high  temperature  of  the  arc  has  practically  no  effect  upon  the  ignitor  surface, 
and  there  is  no  tendency  for  the  mercury  to  wet  the  surface  unless  foreign  material  becomes 
deposited  at  the  contact  area. 

The  ignitor  acts  simply  as  a  resistor;  a  current  flows  through  it  into  the  cathode  pool 
when  voltage  is  applied  between  the  ignitor  terminal  and  the  cathode.  When  sufficient 
current  flows,  an  arc  is  started  between  ignitor  and  pool,  and  conduction  is  established 
through  the  tube  provided  that  positive  anode  voltage  is  applied.  Resistance  of  the  ignitor 
may  vary  from  10  to  100  ohms,  and  ignition  may  occur  at  instantaneous  currents  of  1  or  2 
amp  with  100  to  200  volts  or  at  15  to  30  amp  with  but  a  few  volts  applied.  The  manufac- 
turer's data  should  be  consulted  to  determine  the  exact  ignitor  requirements  to  insure 
proper  operation  throughout  the  life  of  the  tube. 

Formation  of  the  arc  may  be  caused  by  one  or  both  of  two  mechanisms.  At  the  point 
where  the  ignitor  is  immersed  the  mercury  forms  a  meniscus,  so  that  its  surface  is  separated 
from  the  ignitor  surface  by  very  small  distances  at  some  points.  High-voltage  gradients 
exist  across  this  space  and  cause  a  discharge  to  start.  Another  possible  method  of  ignitor 
operation  is  that  of  developing  heat  at  the  several  contact  points  of  the  rough  ignitor 
surface  with  the  mercury  surface.  The  rapid  heating  at  these  sharp  contacts  due  to  the 
ignitor  current  flow  causes  the  mercury  to  vaporize  rapidly  and  break  contact,  thereby 
starting  the  desired  arc. 

Conduction  to  the  anode  occurs  in  the  order  of  20  to  200  microseconds  after  application 
of  the  critical  ignitor  voltage  or  current. 

Ignitor  ratings  include  the  maximum  allowable  peak  voltage  and  peak  current  ratings, 
which  insure  that  the  ignitor  will  not  be  eroded  or  otherwise  affected  during  operation.  A 
maximum  average  current  rating  limits  the  heating  which  may  contribute  to  short  life. 
The  maximum  required  peak  voltage  and  current  specify  the  worst  conditions  of  current 
or  voltage  expected  for  consistent  firing,  and  a  circuit  designed  to  provide  at  least  these 
requirements  will  provide  satisfactory  operation  during  the  tube  life. 

Since  the  arcback  characteristic  of  ignitrons  depends  upon  current,  voltage  and  tem- 
perature, the  tubes  are  rated  for  more  than  one  load  current  depending  upon  the  operating 
voltage  and  duty  cycle.  For  tubes  in  rectifier  service  these  ratings  take  the  form  of  two  or 
more  discrete  current  ratings  at  corresponding  voltages  or  length  of  time  of  operation. 
For  welding  tubes,  a  curve  of  demand  kva  (rms  current  times  rms  circuit  voltage)  vs. 
average  tube  current  is  used,  thus  taking  into  account  the  on  and  off  times  of  the  welder 
as  well  as  the  maximum  current  carried. 

IGNITRON  TESTS.  Because  of  the  size  of  the  tubes  and  equipment  involved  ignitrons 
are  tested  where  possible  in  an  operation  circuit  which  closely  approximates  the  applica- 
tion conditions.  Tests  at  the  maximum  current,  voltage,  and  duty  cycle  should  give  an 
indication,  within  a  very  short  time,  of  the  quality  of  the  tube.  The  arcback  rate  is  fre- 
quently used  as  a  determining  factor  of  this  quality.  The  water-cooling  coils  are  tested 
for  their  capacity  to  withstand  expected  water  pressures,  but  more  important  than  this 
test  may  be  the  user's  test  of  the  purity  of  the  water  being  used,  since  sediment  or  scale  will 
seriously  reduce  heat  transfer  from  the  inner  wall  of  the  tube. 

Ignitors  are  generally  tested  in  a  simple  manner  by  measuring  the  resistance  from  ignitor 
terminal  to  cathode  pool  under  conditions  of  normal  mercury  level.  A  much  more  severe 
test  is  to  apply  a  specified'  firing  voltage  and  measure  the  time  required  for  the  tube  to 
conduct  current,  or  to  measure  both  the  ignitor  current  and  ignitor  voltage  at  which  con- 
duction takes  place. 

EXCITRONS.  The  excitron  is  a  currently  available  type  of  metal  rectifier,  built  in 
both  sealed  and  pumped  units.  This  type  has  the  advantages  of  other  single-anode 
rectifiers  for  production  and  maintenance  and  has  the  firing  characteristics  of  the  multi- 
anode  tanks  in  which  ionization  is  maintained  continuously. 

In  general  construction,  commercially  available  excitrons  are  similar  to  the  sealed  or 
pumped  ignitron  except  that  the  cathode  is  insulated  from  the  tube  wall,  as  it  is  in  the 
metal  mercury-arc  rectifiers.  This  prevents  travel  of  the  arc  to  the  wall  where  the  metal 
may  be  vaporized  and  deposited  over  other  parts  of  the  tube.  In  the  ignitron  type  of 
tube  the  arc  is  extinguished  every  cycle,  and  this  problem  is  not  of  major  importance. 

The  arc  is  established  in  the  excitron  by  a  jet  of  mercury  propelled  up  from  the  cathode 


GENERAL  PHYSICAL  REQUIREMENTS  4-81 

to  contact  an  excitation  anode,  causing  a  temporary  short  circuit  and  arc  formation.  A 
magnetically  controlled  plunger  operates  the  mercury  jet  when  required,  and  the  arc  is 
maintained,  once  established,  by  the  excitation  anode  circuit. 

In  the  use  of  a  control  grid  around  the  anode  and  suitable  baffling  the  excitron  is  similar 
to  ignitrons  and  other  rectifiers. 

BIBLIOGRAPHY 

Hull,  Dr.  A.  W.,  Gas-filled  Thermionic  Tubes,  /.  AJ.E.E.,  VoL  47,  79S-803  (1928). 

Hull,  Dr.  A.  W.,  Hot-cathode  Thyratrons,  Gen.  Elec.  Reo.,  Vol.  32,  213-223,  390-399  (April  and  July 

1929). 

Knowles,  D.  D.,  The  Grid-glow  Tube  Relay,  Electric  J,,  Vol.  25,  176-178  (April  1928). 
Knowles,  D.  D.,  The  Theory  of  the  Grid-glow  Tube,  Electric  J.,  Vol.  27,  116-120  (February  1930). 
Livingston,  O.  W.,  and  H.  T.  Maser,  Shield  Grid  Thyratrons,  Electronics,  Vol.  7,  114-116  (April  1934). 
Morack,  M.  M.,  and  H.  C.  Steiner,  Sealed-tube  Ignitron  Rectifiers,  AJ.E.E.  Trans.,  Vol.  61,  594-598 

(1942). 
Steiner,  H.  C.,  A.  C.  Gable,  and  H.  T.  Maser,  Engineering  Features  of  Gas-filled  Tubes,  Elect.  Eng., 

Vol.  51,  312-317  (May  1932). 
Pike,  0.  W.,  Power  Control  through  Grid-controlled  Heavy  Current  Tubes,  Electronics,  Vol.  1,  85-87 

(May  1930). 
Slepian,  J.,  and  L.  R.  Ludwig,  A  New  Method  of  Starting  an  Arc,  Elect.  Eng.,  Vol.  52,  605-608 

(September  1933). 
Packard,  D.,  and  J.  H.  Hutchings,  Sealed-off  Ignitrons  for  Welding  Control,  AJ.E.E.  Trans..  Vol.  56, 

37-40,  66  (January  1937). 
Germeshausen,  K.  J.,  and  H.  E.  Edgerton,  A  Cold-cathode  Arc-discharge  Tube,  Elect.  Eng*,  July  1936, 

pp.  790  ff. 
Ingram,  S.  B.,  Cold-cathode  Gas-filled  Tubes  as  Circuit  Elements,  AJ.E.E.  Trans.,  Vol.  58,  342  ff 

(1939). 

Arnott,  E.  G.  F-,  Ignitor  Characteristics,  /.  Applied  Phys.,  Vol.  12,  660-669  (September  1941). 
Winograd,  H.,  Development  of  Excitron-type  Rectifier,  AJ.E.E.  Tech.  Paper,  44-78  (March  1944). 


X-RAY  TUBES 

By  S.  Reid  Warren,  Jr. 

22.  GENERAL  PHYSICAL  REQUIREMENTS 

X-ray  tubes  may  be  divided  into  three  general  classes:  (1)  cold-cathode,  gas-filled  tubes; 
(2)  hot-cathode,  high-vacuum  tubes;  (3)  cold-cathode,  high-vacuum  tubes.  The  last- 
named  tube  has  been  used  to  produce  x-ray  exposures  of  short  duration  (approximately 
1  psec)  by  field  emission  (see  reference  8).  Hot-cathode  x-ray  tubes  are  now  used  almost 
exclusively.  Nevertheless  a  description  of  gas-filled  x-ray  tubes  is  of  more  than  historical 
interest,  since  the  phenomena  of  electric  discharge  in  these  tubes  show  clearly  the  limita- 
tions and  the  necessity  for  the  careful  design  of  all  x-ray  tubes. 

The  x-ray  tubes  employed  by  Roentgen  and  his  successors  in  the  field  of  x-ray  research 
up  to  1912  contained  an  anode,  a  cold  cathode,  and  sometimes  an  electrode  called  the 
"  anticathode."  Only  two  electrodes  are 
necessary,  although  it  was  alleged  that 
the  presence  of  the  third  electrode  was 
accompanied  by  greater  stability  of  the 
electric  discharge  through  the  tube. 
They  are  arranged  in  a  glass  bulb  in  the 
manner  shown  in  Fig.  1.  The  tube  is 
exhausted  to  a  pressure  of  a  few  microns 

of  mercury.    The  gas  contains  a  number  _  ^ 

~P  f™~  i~  4.  ~«o  T*  +u«,«  ,v  „  ^^^«+^«i  FIG.  1.  Basic  Elements  of  an  X-ray  Tube  in  Which 
of  free  electrons.  If  there  is  a  potential  Electrons  are  Ejected  from  a  Cold  Cathode  C  by  Ionic 

difference  of  the  order  Of  40  to  80  kv  Bombardment.  G,  the  glass  envelope.  A,  the  anode 
across  the  electrodes  of  the  x-ray  tube,  disk  of  platinum,  tungsten,  or  molybdenum.  <7,  the 
thpsp  fw  AWtrons  arA  ar^lpratpd  in  cath°de,  made  in  the  form  of  a  segment  of  a  sphere  to 
tnese  tree  electrons  are  accelerated  in  focus  the  electrode  stream  upon  a  small  area  of  the 
the  direction  of  the  positive  terminal  or  anode  surface  or  focal  spot, 

anode.    They  acquire  sufficient  velocity 

to  remove  one  or  more  electrons  from  atoms  of  gas  in  their  path.  Tbe  ions  formed  by  the 
ejection  of  these  electrons  from  gas  atoms  have  net  positive  charges.  Under  the  action  of 
the  electric  field  these  positive  ions  are  accelerated  toward  the  cathode.  When  the  posi- 
tive ions  impinge  upon  the  cathode,  electrons  are  ejected  from  it.  They  in  turn  are  at- 
tracted toward  the  anode  and  acquire  a  velocity  depending  upon  the  anode-cathode  volt- 
age. The  electrons  which  impinge  upon  the  anode  at  high  velocity  cause  the  generation 
of  x-rays  near  the  surface  of  the  anode.  As  a  result  of  the  positive-ion  stream  passing 
toward  the  cathode  and  the  electron  stream  passing  to  the  anode,  a  current  flows  through 


4-82  ELECTRON  TUBES 

the  x-ray  tube,  which,  in  accordance  with  convention  concerning  the  direction  of  current 
flow,  is  said  to  pass  from  anode  to  cathode. 

The  process  involved  in  the  generation  of  x-rays  at  voltages  of  the  order  of  100  kv  is 
extremely  inefficient.  Of  the  total  electrical  energy  supplied  to  an  x-ray  tube  only  about 
0.01  per  cent  is  transformed  into  useful  x-rays.  Of  the  remainder,  the  greatest  part  is 
dissipated  as  heat  from  the  x-ray  tube  anode. 

The  current  through  an  x-ray  tube  of  this  type  cannot  be  controlled  independently  of 
the  voltage.  As  the  parts  become  heated  during  operation  the  gas  pressure  within  the 
tube  changes,  often  in  an  unpredictable  manner,  apparently  influenced  as  much  by  the 
previous  history  of  the  tube  as  by  the  contemporary  electrical  input  conditions.  If  the 
pressure  within  the  tube  decreases,  a  higher  anode-cathode  voltage  is  required  to  cause  the 
flow  of  a  given  current.  The  x-rays  generated  at  high  voltage  with  low  gas  pressures  are 
more  penetrating  than  those  generated  at  lower  voltages  with  higher  gas  pressures.  Low- 
pressure,  cold-cathode  tubes  and  the  highly  penetrating  x-rays  generated  by  operating 
them  at  high  voltages  are  called  hard  tubes  and  hard  x-rays.  Tubes  with  relatively  high 
pressure  operated  at  low  voltages  are  called  soft,  and  the  radiation  so  generated  is  called 
soft  radiation.  The  terms  hard  and  soft  are  purely  qualitative. 

In  spite  of  ingenious  attempts  to  make  a  practically  useful  cold-cathode  x-ray  tube,  it 
has  been  used  very  little  since  the  invention  in  1913  of  the  hot-cathode  x-ray  tube  by 
W.  D.  Coolidge  and  his  co-workers. 

Two  important  investigations  during  the  period  1908-1913  contributed  to  the  develop- 
ment of  an  x-ray  tube  in  which  the  current  can  be  varied  independently  of  the  voltage. 
Langmuir  had  investigated  thoroughly  the  electronic  emission  from  hot  metallic  filaments 
noted  by  Edison  and  earlier  by  Elster  and  Geitel.  Briefly,  to  summarize  this  work,  the 
following  important  findings  are  noted: 

1  .  Unless  the  pressure  of  the  gas  within  the  tube  is  less  than  about  1  micron  of  mercury, 
positive  ions  produced  in  the  same  manner  as  those  in  the  gas  x-ray  tube  described  above 
cause  an  important  variation  in  the  electron  output  of  the  filaments.  At  pressures  lower 
than  this  the  electrons  emitted  by  the  hot  body  are  due  apparently  entirely  to  thermionic 
emission,  and  therefore  their  number  depends  only  upon  the  cathode  material  and  its 
temperature. 

2.  In  general  the  introduction  of  various  gases  causes  a  considerable  decrease  in  electron 
emission,  particularly  at  low  filament  temperatures.    This  effect  should  not  be  confused 
with  the  effects  due  to  ions  formed  in  the  added  gas. 

3.  At  very  low  gas  pressures  the  thermionic  current  from  a  tungsten  filament  varies  with 
the  temperature  in  the  following  manner: 


T  is  the  absolute  temperature;  i,  the  current  in  amperes  per  square  meter  of  cathode 
surface;  for  tungsten  a  =  6.02  X  105  amp  per  sq  m  per  (degree)  2;  b  =  52,400  deg  K.  An 
expression  similar  to  this  was  first  developed  by  Richardson. 

4.  The  operation  of  a  tube  with  high  vacuum  is  greatly  influenced  by  a  space  charge 
consisting  of  a  cloud  of  electrons  surrounding  the  hot  filament. 

Investigators  have  found  that  x-rays  are  generated  more  efficiently  by  elements  with 
high  atomic  number.  Therefore  they  attempted  to  find  a  metal  with  the  following  charac- 
teristics: 

1.  High  atomic  number,  to  provide  efficient  x-ray  generation. 

2.  High  heat  conductivity,  to  allow  for  dissipation  of  the  heat  generated  at  the  x-ray 
tube  anode. 

3.  Low  vapor  pressure,  to  assure  stable  operation. 

4.  High  melting  point,  to  provide  sufficient  capacity. 

5.  Ease  of  machining  and  relatively  low  cost. 

X-ray  tubes  employed  by  workers  who  immediately  followed  Roentgen  contained 
anodes  of  platinum.  It  was  found  subsequently  that  tungsten  was  better  suited  than 
platinum  in  all  respects  except  that  it  was  impossible,  by  means  developed  up  to  that  time, 
to  purify  and  to  machine.  A  period  of  several  years  was  required  to  devise  methods  for 
producing  and  working  tungsten.  The  manufacture  of  uniform,  standard  x-ray  tubes 
depends  upon  the  carefully  controlled  processing  of  the  component  metal  and  glass  parts. 
The  method  now  generally  used  requires  the  removal  and  purification  of  an  oxide  of  tung- 
sten from  the  ore  wolframite.  The  powdered  oxide  is  formed  into  blocks  and  heated  in 
the  presence  of  hydrogen.  After  such  treatment,  the  oxide  is  reduced  to  metallic  tungsten; 
the  bar  contracts  but  remains  extremely  weak  mechanically.  It  is,  however,  sufficiently 
strong  so  that  bars  about  2  cm  square  and  15  to  20  cm  long  may  be  supported  at  one  end 
by  a  water-cooled  copper  electrode,  while  the  other  end  rests  in  a  pool  of  mercury  which  is 
also  cooled.  This  system  is  surrounded  by  a  jar  through  which  hydrogen  passes.  An  elec- 


TUBES  FOE  X-RAY  THERAPY 


4-83 


trie  current  is  passed  through  the  tungsten  bar,  heating  it  to  about  2800  deg  cent.  Gradu- 
ally the  metal  contracts  and  becomes  mechanically  stronger.  It  is  then  heated  to  red  heat 
in  air  and  swaged.  This  process  consists  of  operating  upon  the  tungsten  bar  with  tools 
which  pound  it  radially  from  several  directions.  By  this  treatment  the  tungsten  becomes 
hard,  homogeneous,  and  ductile.  It  is  possible  to  continue  the  swaging  process  until  the 
bar  has  been  reduced  to  a  diameter  of  a  few  millimeters.  It  may  then  be  drawn  into  tung- 
sten wire  by  means  of  diamond  dies.  The  larger  bars  may  be  cut  or  polished  by  means  of 
Carborundum.  Solid  tungsten  anodes  for  x-ray  tubes  may  be  constructed  from  such  bars. 
Smaller  sections  can  be  cut  by  means  of 
thin  (0.01  in.)  rubber  disks  a  few  inches 
in  diameter  containing  240-grit  Carbo- 
rundum and  operating,  wet,  at  high 
speeds  (9000  rpm). 

W.  D.  Coolidge  devised  an  x-ray  tube 
like  the  one  shown  in  Fig.  2.  He  found 
that  it  had  the  following  characteristics: 

1.  The  tube  allows  current  to  pass 
only  in  the   direction  from  anode  to 
cathode. 

2.  The    current   through   the    x-ray 
tube  is  practically  independent  of  the 
voltage  applied  to  the  anode  and  cathode 

for  voltages  of  30  to  200  kvp.    The  current  may  be  varied  by  changing  the  temperature 
of  the  tungsten  cathode. 

3.  The  area  upon  the  anode  over  which  electrons  impinge  may  be  controlled  by  the 
catfcode  shield  S  and  by  the  position  of  the  cathode  coil  C  within  this  shield. 

X-ray  tubes  are  employed  for  the  following  purposes: 

VOLTAGE  CTTKBENT 

PUBPOSE  LIMITS  LIMITS 

1.  Generation  of  x-rays  for  the  treatment  of  disease 

(x-ray  therapy) 12-2000  kvp  0.1-  30  ma 

2.  The  production  of  x-ray  shadow  pictures  for  medical 

diagnosis  (roentgenography) 30-  110  kvp         10     —500  ma 

3.  Industrial  roentgenography 30-2000  kvp  0. 1-100  ma 

4.  X-ray  diffraction 10-  100  kvp  2     -100  ma 

Tubes  for  these  various  purposes  differ  considerably  in  structure.    Some  of  the  tubes 
now  available  are  described  in  the  following  sections. 


FIG.  2.  JBasic  .Clements  oi  an  JL-ray  Tube  in  Wi 
Electrons  are  Ejected  from  a  Hot  Cathode  Cby  Therm- 
ionic Action.  (?,  the  glass  envelope.  A,  the  anode  disk, 
usually  of  tungsten.  C,  the  cathode  coil  of  tungsten 
wire.  S,  the  metal  focusing  shield  connected  electrically 
to  one  of  the  cathode  leads. 


23.  TUBES  FOR  X-RAY  THERAPY 

Roentgenologists  require,  for  general  x-ray  therapy,  tubes  that  will  operate  under  the 
following  conditions: 

1.  Tube  voltages  of  50  to  400  kvp  and  tube  currents  of  2  to  30  ma  (average),  supplied 
by  constant-potential,  full-wave  pulsating,  or  half-wave  pulsating  high-voltage  generators. 

2.  Exposures  of  0.5  to  30  Tnin  duration  are  used. 

3.  Insulation  to  rninirnize  the  hazard  of  electrical  shocks  to  patient  and  operator  are 
essential. 

4.  X-rays  must  be  confined  to  a  cone  the  axis  of  which  passes  through  the  tube  target 
and  the  part  of  the  patient's  body  to  be  treated,  to  minimise  the  possibility  of  x-ray  burn 
to  the  operator  and  patient. 

The  length  of  an  x-ray  tube  depends  upon  the  maximum  anode-cathode  voltage  to 
which  it  may  be  subjected.  The  ends  of  the  tube,  to  which  the  electrical  connections  are 
made,  must  be  separated  sufficiently  to  prevent  sparkover.  The  electrical  stress  in  the 
glass  envelope  must  be  sufficiently  low  to  prevent  puncture.  The  tube  must  be  mechani- 
cally strong  enough  to  support  the  anode  and  cathode  assemblies  rigidly.  Therapy  tubes, 
for  use  at  50  to  400  kvp,  immersed  in  oil,  are  0.2  to  1.5  m  long.  Formerly  the  glass  en- 
velopes were  made  of  soft  glass.  Manuf acturing  methods  devised  recently  have  led  to  the 
use  of  a  hard  glass  such  as  Pyrex.  Pyrex  has  superior  heat-resisting  qualities,  greater 
dielectric  strength,  and  greater  mechanical  strength  than  soft  glass.  Air-cooled  therapy 
tubes  have  a  spherical  glass  bulb,  15  to  25  cm  in  diameter,  surrounding  the  anode  and 
cathode.  This  method  of  construction  has  the  following  advantages: 

1.  The  sphere  radiates  more  heat  than  a  small  cylindrical  envelope. 

2.  The  glass  is  far  enough  from  the  anode  and  cathode  so  that  it  is  not  stressed  by  strong 
electric  fields  caused  by  the  applied  anode-cathode  voltage. 


4-84 


ELECTRON  TUBES 


3.  Stray  electrons  do  not  collect  on  the  interior  of  the  spherical  surface  in  sufficient 
quantity  to  cause  unstable  operation  or  puncture  of  the  tube. 

In  the  so-called  Universal  tube,  Fig.  3,  a  bar  of  molybdenum  0.5  cm  in  diameter  is 
supported  by  glass  at  the  end  of  the  anode  stem  and  extends  into  the  tube,  concentric 
with  its  axis,  to  a  point  about  5  cm  from  the  tube  center.  At  this  end  of  the  bar  a  solid 

tungsten  block  is  fastened.  This  block  is 
truncated  at  an  angle  of  45°  with  the  tube 
axis.  The  minor  axis  of  the  elliptical  face  thus 
formed  is  2  cm  long.  The  minor  axis  of  the 
target  or  focal  spot  is  approximately  1.8  cm. 

Fm.3.     UnHersal  TherapylTube,  Solid  Tung-    Heat  _  is   dissipated   chiefly   by  radiation     A 
sten  Anode.    Air-cooled.  spherical  glass  bulb  is  required  to  aid  this  heat 

dissipation. 

The  tubes  with  spherical  bulbs  are  difficult  to  mount  and  to  shield  by  reason  of  their 
size.  Modern  therapy  tubes  are  therefore  designed  for  operation  immersed  in  oil,  with 
cooling  of  the  anode  by  means  of  oil  that  is  forced  against  the  back  of  the  anode  by  a 
pump.  Special  design  of  the  electrodes,  the  use  of  hard-glass  envelopes,  and  oil  immersion 
result  in  tubes  that  are  considerably  smaller  than  tubes  designed  for  mounting  in  air.  An 
example  of  such  a  tube  is  shown  in  Figs.  4  and  5,  designed  to  operate,  in  oil,  at  250  kvp, 
15  ma.  Note  the  hollow  cylindrical  projection  (hood)  from  the  region  of  the  tungsten 
target  toward  the  cathode;  this  hood  prevents  secondary  electrons  from  impinging  on  the 
inner  surface  of  the  glass  envelope. 

The  cathode  coil  is  usually  a  tungsten  spiral  which  operates  at  5  to  10  volts,  3  to  6  amp. 
The  coil  is  supported  and  surrounded  by  a  metal  cup  to  shield  the  glass  from  secondary 
electrons  and  to  focus  the  electron  stream  upon  the  target  or  focal 
spot  of  the  anode.  In  a  therapy  tube  a  small  focal  spot  or  source 
of  x-rays  is  not  necessary.  The  focusing  action  of  the  cup  is 
limited  to  preventing  electrons  from  going  past  the  anode  face 
into  the  anode  stem. 

Tubes,  such  as  that  shown  in  Figs.  4  and  5,  are  often  placed  in 
a  grounded  metal  tube  head,  supported  by  means  of  a  tube  stand 
that  permits  easy  mechanical  manipulation  of  the  tube.  High- 
voltage  leads  are  flexible  shockproof  cables  with  a  grounded  ex- 
ternal sheath.  Figure  6  shows  an  apparatus  of  this  kind  being 
adjusted  to  make  an  x-ray  film  of  a  weld;  therapy  tube  stands  are 
essentially  similar.  The  tube  head  and  the  rectangular  cone  ex- 
tending from  the  head  toward  the  weld  are  lead  lined;  thus  the 
x-ray  output  is  confined  to  a  relatively  sharply  defined  beam 
through  the  cone. 

All  x-ray  tubes  used  for  the  routine  treatment  of  deep-seated 
tumors  are  evacuated  to  a  pressure  of  10  "^  micron  of  mercury  or 


\ 


Steel 
Cathode 


Copper  Hooded 
Anode 

Nickel  Cooling 
Coil 


FIG.  4.  X-ray  Tube 
Designed  for  Opera- 
tion, Immersed  in 
Oil,  at  250  kvp,  15 
ma,  Using  Forced- 
oil  Cooling.  (Cour- 
tesy General  Elec- 
tric X-ray  Corpora- 
tion.) 


Tungsten  ,,,,,, 

Window    Ground  Window  In 
BorosiUcate  Glass 

FIG.  5.    Axial  Section  of  Tube  Shown  in  Fig.  4. 
General  Electric  X-ray  Corporation.) 


(Courtesy 


less.  It  is  necessary  not  only  to  evacuate  to  this  pressure  while  the  anode  and  cathode 
are  cold  but  also  to  heat  the  metal  parts  by  means  of  high-frequency  induction  while  the 
pumping  proceeds.  Before  being  sealed  off,  the  tube  is  operated  at  an  anode-cathode 
voltage  and  current  slightly  in  excess  of  the  tube  rating  to  insure  thorough  degassing  of 
the  metal  parts. 


TUBES  FOR  X-RAY  THERAPY 


4-85 


HIGH-VOLTAGE    THERAPY    TUBES.      X-ray 

apparatus  for  therapy  and  for  industrial  radiography 
at  tube  voltages  of  1000  and  2000  kvp  are  contained 
in  steel  tanks,  in  which  air  or  "Freon"  is  maintained 
at  a  pressure  equivalent  to  several  atmospheres.  The 
source  of  high-voltage,  mounted  in  the  tank  with  the 
x-ray  tube,  is  a  van  de  GraafT  generator  or  a  specially 
designed  transformer  operating  at  180  cycles  per 
second.  The  tubes  are  sealed,  with  the  anode  placed 
at  one  end  of  the  cylinder;  the  cathode  is  placed  at 
the  other  end.  The  anode  is  grounded,  and  the  focal 


FIG.  6.     220-kvp  Shockproof  X-ray  Tubehead  and  Tube- 
stand.      (Courtesy  X-ray  Division,  Westinghouse  Electric 
Corporation.) 

spot  is  cooled  by  water  that  flows  over  the  end  of  the 
tube.  A  tube  rated  at  2000  kvp  is  shown  in  Fig.  7. 
This  tube  has  a  gold  target;  there  is  some  evidence 
that  the  surface  of  the  focal  spot  is  molten  during 
operation,  although  there  is  no  evidence  of  the  in- 
stability that  might  be  expected  to  accompany  this 
condition.  Note  that  the  tube  is  equipped  with  a 
series  (180)  of  accelerating  anodes.  These  are  re- 
quired to  maintain  uniform  potential  gradient  along 
the  tube  and  to  minimize  the  effects  of  electrons' 
diverging  from  the  beam.  The  magnetic  focusing 
coil  shown  in  Fig.  7  insures  stable  size,  shape,  and 
position  of  the  focal  spot  on  the  gold  target.  The 
tube  is  approximately  9  ft  long;  the  extension  from 
the  lowest  accelerating  anode  to  the  target  is  3  ft. 
The  rating  is  2000  kvp,  300  juamp.;  the  focal  spot, 
with  magnetic  focusing,  is  approximately  2.5  mrn  in 
diameter. 


2^-Cathode 


Accelerating 
Electrodes 


.Focusing 
Magnet 


Water 
-Cooling 


Gold 
/target 


FIG.  7.     Tube  for  Therapy  and  In- 
dustrial     Radiography;      Operating 
Voltage  2000  kvp.    (Courtesy  Mach- 
lett  Laboratories.) 


4-86  ELECTRON  TUBES 

24.  TUBES  FOR  MEDICAL  ROENTGENOGRAPHY  AND 
ROENTGENOSCOPY 

To  produce  diagnostically  useful  roentgenograms  of  parts  of  the  human  body,  x-ray 
tubes  with  the  following  characteristics  are  required: 

1.  The  focal  spot  or  source  of  x-rays  must  be  as  small  as  possible  in  order  to  obtain 
sharp  shadow  borders. 

2.  The  technique  of  roentgenography  prescribes  x-ray  tube  voltages  of  30  to  110  kvp, 
x-ray  tube  currents  of  10  to  500  ma  (average),  and  exposure  times  of  1/30  to  30  sec. 

3.  The  primary  x-ray  beam  must  be  confined  to  a  cone  with  the  focal  spot  at  the  apex 
and  the  film  at  the  base  to  protect  the  operator  from  x-ray  burn. 

4.  Insulation  of  the  tube  and  high-voltage  leads  is  advisable  but  not  necessary  (except 
in  dental  roentgenography)  since  the  focal  spot-film  distances  generally  used  are  greater 
than  60  cm. 

Except  for  the  requirement  that  the  focal  spot  be  small,  the  same  principles  are  applied 
in  designing  roentgenographic  tubes  as  those  described  above  for  therapy  tubes.  The 
need  for  minimizing  the  focal  spot  size  having  been  prescribed,  certain  practical  factors 
are  found  to  oppose  this  decrease: 

1.  For  any  particular  method  of  anode  construction  the  energy  dissipated  at  the  focal 
spot  during  an  exposure  cannot  exceed  the  value  which  will  melt  the  tungsten  target. 
The  melting  point  of  tungsten  is  3370  deg  cent. 

2.  Owing  to  the  thickness  of  the  part  to  be  roentgenographed,  the  shadows  of  parts 
farthest  from  the  film  are  more  distorted  than  those  near  the  film.    The  focal  spot-film 
distance  must  be  increased  to  decrease  this  effect.    The  energy  supplied  to  the  tube  in- 
creases in  proportion  to  the  square  of  the  focal  spot-film  distance.    Therefore,  the  focal 
spot  area  must  be  increased  in  like  proportion.    In  other  words,  if  the  focal  spot-film  dis- 
tance is  increased  to  reduce  the  magnification  of  shadows  of  parts  at  a  distance  from  the 
film,  the  focal  spot  must  be  enlarged,  the  energy  supplied  to  the  tube  must  be  increased, 
and  the  sharpness  of  the  shadows  remains  unchanged. 

3.  Parts  of  the  body  are  continually  in  motion.    Roentgenograms  of  such  parts  must  be 
made  in  as  short  a  time  as  possible  to  minimize  the  unsharpness  or  blurring  caused  by 
motion  during  exposure.    The  minimization  of  exposure  time  requires  an  increase  in  focal 
spot  size,  all  other  factors  remaining  constant.    For  example,  roentgenogram  of  the  chest 
of  a  cadaver  may  be  made  with  a  tube  having  a  focal  spot  less  than  1  mm  square.    To 
make  a  roentgenogram  of  the  chest  of  a  normal  adult  requires  a  tube  with  a  focal  spot  at 
least  3  mm  square.    Even  this  is  a  compromise,  for  no  matter  how  short  the  exposure  time 
some  unsharpness  is  produced  by  motion. 

Nearly  all  roentgenographic  tubes  now  manufactured  are  constructed  and  used  so  that 
the  line  from  the  focal  spot  to  the  center  of  the  film  makes  an  angle  of  10°  to  45°  with  the 
anode  face.  The  projection  of  the  actual  focal  spot  (the  source  of  x-rays)  upon  a  particular 
point  in  the  film  is  called  the  effective  focal  spot  area  at  that  point.  Figure  8  illustrates 
three  types  commonly  used.  The  effective  focal  spot  Fz  for  the  45°  tube  is  a  circle  6  mm 
in  diameter.  The  effective  focal  spots  for  the  20°  and  10°  tubes  having  the  same  target 
areas  as  the  45°  tube  are  squares  with  sides  of  3.7  mm  and  2.6  mm,  respectively.  The  max- 
imum allowable  exposure  energies  (for  exposure  times  less  than  0.2  sec)  are  equal  for  the 
three  tubes,  since  the  actual  focal  spot  areas  are  equal.  Therefore,  for  a  given  maximum 
rating  the  sharpness  of  the  roentgenographic  image  increases  as  the  angle  between  the 
focal  spot  and  central  x-ray  beam  decreases.  Two  limitations  prescribe  the  minimum 
focal  spot-film  distances  at  which  a  tube  may  be  used: 

1.  The  intensity  of  x-rays  emanating  from  the  focal  spot  within  a  few  degrees  of  the 
tangent  to  the  anode  face  is  less  than  the  intensity  of  the  central  beam  at  the  same  dis- 
tance from  the  tube.    The  intensity  must  be  practically  uniform  over  the  area  of  the  x-ray 
film,  in  the  absence  of  absorbing  objects  between  tube  and  film. 

2.  The  effective  focal  spot  size  and  the  distortion  vary  over  the  film  area.    This  varia- 
tion must  be  minimized  to  insure  that  the  roentgenogram  is  diagnostically  useful  over  its 
entire  area. 

The  design  of  roentgenographic  tubes  is  determined  in  general  by  the  criteria  described 
above  in  article  23,  "Tubes  for  X-ray  Therapy."  There  are  three  requirements  which 
differ  qualitatively: 

1.  The  anode-cathode  voltages  are  less  for  roentgenography  than  for  therapy. 

2.  It  is  extremely  important  to  produce  a  small,  uniform  effective  focal  spot  in  roent- 
genographic tubes. 

3.  Exposure  times  for  roentgenography  are  relatively  short,  requiring  that  the  anode 
have  a  high  heat  capacity. 


TUBES  FOR  MEDICAL  ROENTGENOGRAPHY 


4-87 


Modern  roentgenographic  tubes  use  the  line  or  band  focus  shown  in  the  20°  and  10° 
sketches,  Fig.  8.  The  target  is  a  disk  of  tungsten  about  1.5  mm  thick,  usually  with  an 
area  two  or  three  times  the  actual  focal  spot  area.  The  tungsten  is  embedded  in  a  heavy 
copper  bar.  Suitable  thermal  contact  between  the  tungsten  and  copper  is  difficult  to 
obtain.  The  usual  procedure  involves  fixing  the  tungsten  disk  at  the  bottom  of  a  graphite 
crucible  and  pouring  molten  electrolytic  copper  over  the  disk.  The  casting  is  accomplished 


A— Anode 
B=X-ray  Film 
Fi=ActuaI  Focal  Spot 
F2=  Effective  Focal  Spot 


•1 

FIG.  8.    Relations  of  Actual  Focal  Spots,  FI,  and  Effective  Focal  Spots,  F%,  for  Roentgenographic  Tube 
Anodes.    Actual  focal  spot  areas  of  all  three  tubes  are  equal. 

in  an  evacuated  chamber  to  prevent  the  formation  of  any  compounds  that  might  prohibit 
the  adherence  of  the  copper  to  the  tungsten. 

With  the  type  of  construction  just  described,  the  maximum  allowable  exposure  rating 
for  exposures  of  1  sec  or  less  is  250  watts  per  square  millimeter  of  actual  focal  spot  area. 
For  long  exposures  the  rating  depends  upon  the  efficiency  of  the  heat-radiating  system, 
and  therefore  it  is  specified  for  each  type  of  tube  by  the  manufacturer.  These  data  are 
given  in  graphical  form,  called  tube  rating  charts. 

Shock-proof  roentgenographic  tubes  are  constructed  similarly  to  the  therapy  tube 
shown  in  Fig.  6. 

To  increase  the  rating  for  a  focal  spot  of  a  given  size,  tubes  have  been  developed  with 
rotating  anodes,  the  cathode  stream  being  so  directed  that  the  focal  spot  is  separated 
several  millimeters  from  the  tube  axis.  Different  areas  of  the  rotating  anode  therefore  act 
as  focal  spot  during  the  exposure.  Two  such  tubes  are  shown  in  Figs.  9-12.  The  anode 


FIG.  9.  Rotating-anode  Roentgenographic  Tube.  Ratings  (in  oil-insulated  tubehead):  2-ram  focal 
spot,  maximum  voltage  100  kvp,  500  ma  at  90  kvp  for  1/30  sec,  400  ma  at  80  kvp  for  1/5  sec;  1-mm  focal 
spot,  maximum  voltage  100  kvp,  200  ma  at  90  kvp  for  1/20  sec,  100  ma  at  90  kvp  for  3  sec.  (Courtesy 

Machlett  Laboratories.) 

within  the  evacuated  tube  envelope  operates  as  the  rotor  of  a  split-phase  induction  motor. 
These  tubes  are  equipped  with  two  cathodes;  one  produces  an  effective  focal  spot  1mm 
square;  the  other,  2  mm  square.  The  rotors  operate  at  approximately  3000  rpm. 

The  tube  shown  in  Figs.  9  and  10  is  equipped  with  an  anode  comprising  a  tungsten  disk 
approximately  6  cm  in  diameter  and  2.3  mm  thick.  This  disk  is  mechanically  connected 
to  the  rotor  of  the  driving  motor  by  means  of  a  short  molybdenum  shaft;  the  entire  assem- 


ELECTRON  TUBES 


bly  is  blackened  to  promote  radiation  of  heat  to  the  oil  surrounding  the  tube  when  it  is 
operating  in  its  shockproof  tube  head.    The  balls  of  the  bearing  within  the  vacuum  tube 

are  coated  with  a  thin,  uniform  film  of  silver,  which 
reduces  bearing  friction.  Samples  of  the  manufac- 
turer's ratings  are  listed  below  Fig.  9. 

The  anode  structure  of  the  tube  shown  in  Figs.  11 
and  12  consists  of  a  tungsten  cap  backed  by  a  heavy 
copper  blackened  cylinder,  the  purpose  of  which  is  to 
provide  high  heat  storage  and  a  large  area  for  the 
radiation  of  heat  to  the  oil  surrounding  the  tube  en- 
velope. The  ball  bearings  of  this  tube  are  lubricated 
by  thin  films  of  barium. 


FIG.  10.    Section  of  Rotating  Anode 

Tube  Shown,  in  Fig.  9.     (Courtesy 

Machlett  Laboratories.) 


FIG.  11.  Rotating-anode  Roentgenographic  Tube. 
Ratings  (in  oil-insulated  tubehead):  2-mm  focal 
spot,  maximum  voltage  100  kvp,  500  ma  at  90  kvp 
for  1/30  sec,  400  ma  at  80  kvp  for  2/5  sec;  1-mm  focal 
spot,  maximum  voltage  100  kvp,  200  ma  at  94  kvp 
for  1/20  sec,  100  ma  at  90  kvp  for  5.6  sec.  (Cour- 
tesy General  Electric  X-ray  Corporation.) 


Borosilicate 
Glass 


Insulating 
Shield 


Stator 


Rotating-anode  tubes  are  usually  equipped  with  counters  to  record  the  number  of 
exposures,  and  with  control  circuits  that  prevent  high-voltage  excitation  of  the  tube  unless 

the  anode  is  rotating  at  rated  speed 
(approximately  3000  rpm) . 

Tubes  used  for  roentgenoscopy  (view- 
ing of  x-ray  shadows  by  means  of  a 
fluorescent  screen)  are  similar  to  the 
stationary-anode  tubes  described  above, 
except  that  focal  spots  less  than  2  mm 
square  may  be  used,  since  the  current 
seldom  exceeds  10  ma.  These  tubes  are 
often  non-shockproof,  since  they  are 
commonly  mounted  under  a  fluoroscopic 
table,  out  of  reach  of  operator  and 
patient. 

Modern  roentgenographic  and  fluoro- 
scopic tubes  should  be  used  about  10-20 


Steel 


Tungsten 


Copper  Anode  and 
Rotor  Housing 

Blackened  Surface 

FIG.  12.    Section  of  the  Rotating-anode  Tube  Shown 
in  Fig.  11.    (Courtesy  General  Electric  X-ray  Corp- 
oration.) 


per  cent  below  the  manufacturers' 
ratings  to  guarantee  a  long  useful  life. 
Roentgenographic  tubes,  so  operated, 
often  can  be  used  for  100,000  exposures 
or  more. 


BIBLIOGRAPHY 


4-89 


25.  TUBES  FOR  INDUSTRIAL  ROENTGENOGRAPHY 
AND  FLUOROSCOPY,  AND  FOR  X-RAY  DIFFRACTION 

The  use  of  x-rays  (and  of  gamma  rays)  for  the  examination  of  commercial  products  has 
greatly  increased,  particularly  during  World  War  II;  such  examinations  are  quick  and 
non-destructive.  The  following  are  representative  of  the  diverse  applications  of  this 
method  of  inspection:  the  examination  for  flaws  in  castings  and  welded  joints;  the  inspec- 
tion of  packaged  foods  for  foreign  bodies;  the  final  checking  of  manufacturing  tolerances 
in  such  items  as  golf  balls  and  multielectrode  vacuum  tubes;  the  inspection  of  packages  for 
contraband;  the  inspection  of  citrus  fruits  suspected  of  being  damaged  by  frost;  the  in- 
spection of  paintings  and  mummies. 

X-ray  tubes  for  industrial  radiography  and  fluoroscopy  are  similar  to  those  described 
in  articles  23  and  24.  The  desire  to  use  x-ray  tube  voltages  up  to  2  Mv  for  radiography  of 
thick  (as  much  as  0.3-0.4  m)  steel  was,  in  fact,  a  major  stimulus  to  the  development  of 
the  high-voltage  tubes  described  in 
article  23  (Fig.  7) .  In  all  industrial 
radiographic  tubes,  regardless  of 
voltage,  as  small  a  focal  spot  as 
possible  must  be  produced  to  mini- 
mize the  unsharpness  of  shadow 
borders,  just  as  in  medical  roent- 
genography. 


FIG.  13.    Shockproof  X-ray  Diffraction  Tube,  with  Grounded, 

Water-cooled  Anode  of  Molybdenum,  Copper,  Cohalt,  or 

Iron,  and  a  Window  of  Beryllium.    Eatings  50  kvp,  10-20 

ma,  continuous.    (Courtesy  Machlett  Laboratories.) 


Since  industrial  radiographic  equipment  may  have  to  be  moved  about  in  a  factory  to 
use  it  near  a  heavy  object  to  be  inspected,  design  of  equipment  is  based  upon  movabUity, 
flexibility  of  adjustment,  and  adequate  protection  of  personnel  from  electric  shock  and 
excessive  exposure  to  x-rays.  Thus  all  industrial  radiographic  apparatus  now  manufac- 
tured is  shockproof  and  shielded  with  lead  or  other  x-ray-absorbing  material,  except  in 
the  direction  of  the  useful  beam. 

The  investigation  of  the  structure  of  crystalline  materials  by  means  of  x-ray  diffraction 
has  been  developed  from  the  early  (1908-1915)  theoretical  analysis  and  experiments  of 
von  Laue,  Friedrich,  Knipping,  W,  H.  and  W.  L.  Bragg,  and  their  colleagues.  In  recent 
years,  x-ray  diffraction  apparatus  suitable  for  routine  analyses  of  crystal  structure  has 
been  used  in  industry.  X-ray  tubes  for  this  purpose  consist  of  an  anode  of  tungsten, 
molybdenum,  copper,  iron,  or  cobalt,  a  conventional  incandescent  tungsten  cathode,  and 
one  or  more  windows  of  beryllium,  which  has  an  extremely  low  x-ray  absorption  coefficient. 


FIG.  14.     Section  of  X-ray  Diffraction  Tube  Shown  in  Fig.  13.     (Courtesy  Machlett  Laboratories.) 

The  tubes  operate  at  10-50  kvp,  10-50  ma.  A  selective  filter  may  be  used  to  isolate  the 
characteristic  radiation  from  the  anode,  thereby  producing  a  relatively  intense,  almost 
monochromatic  beam  of  x-rays.  A  photograph  of  an  x-ray  diffraction  tube  and  a  cross- 
Bection  of  the  tube  are  shown  in  Figs.  13  and  14. 


BIBLIOGRAPHY 

1.  Kaye,  G.  W.  C.,  X-rays,  Longmans,  Green  (1926). 

2.  Coolidge,  W.  D.,  A  Powerful  Roentgen-Ray  Tube  with  a  Pure  Electron  Discharge,  Phys.  Rev., 

Vol.  2,  409  (1913). 

3.  Langmuir,  Irving,  The  Effect  of  Space  Charge  and  Residual  Gases  on  Thermionic  Currents  in 

High  Vacuum,  Phys.  Rev.,  Vol.  2,  No.  6  (December  1913). 

4.  Bouwers,  A.,  An  X-ray  Tube  with  Rotating  Anode,  Physics,  Vol.  10,  125  (1930). 

5.  St.  John,  A.,  and  H.  R.  Isenburger,  Industrial  Radiography,  2nd  Ed.     John  Wiley  (1943). 

6.  Siegbahn,  M.,  The  Spectroscopy  of  X-rays.    Oxford  University  Press  (1925). 

7.  Clark,  G.  L.,  Applied  X-rays,  3d  Ed.    McGraw-Hill  (1940). 

8.  Slack,  C.  M,,  and  L.  F.  Ehrke,  Field  Emission  X-ray  Tube,  J.  Applied  Phys.,  Vol.  12,  165-168 

(February  1941). 

9.  Machlett,  R.  R.,  and  T.  H.  Rogers,  A  New  Design  for  Increasing  the  Heat-dissipating  Capacity 

of  Rotating  Anode  Tubes,  Am.  J.  Roentgenology  &  Radium  Therapy,  Vol.  48,  685-690  (November 
1942). 

10.  Atlee,  Z.  J.,  Design  and  Application  of  X-ray  Tubes,  Electronics,  Vol.  13,  26-30,  62,  64  (October 
1940). 


SECTION  5 
ELECTRIC  CIRCUITS,  LINES,  AND  FIELDS 


THEORY  OF  LINEAR  PASSIVE  NETWORKS 


ART. 


BT  p-  H-  RICHARDSON 


PAGE 


1.  Non-sinusoidal  Currents  and  Voltages.  .  02 

2.  Single-mesh  Circuit  ..................  03 

3.  The  Complex  Frequency  Plane  ........  04 

4.  Mesh  Equations  .....................  05 

5.  Nodal  Equations  .....................  06 

6.  Two-terminal  Impedances  .............  07 

7.  Four-terminal  Networks  ..............  10 

8.  Power  Transfer  ......................  15 

9.  Distortion  ...........................  16 

10.  Corrective  Networks  .................  16 

RECURRENT  NETWORKS 
BT  P.  H.  RICHARDSON 

11.  Symmetrical  Networks  ..............  23 

12.  Uniform  Lines  —  Networks  with  Distrib- 

uted Constants  ....................  24 

TRANSIENTS  IN  NETWORKS 
BY  HAROLD  A.  WHEELER 

13.  Transient  Disturbances  ...............  26 

14.  Behavior  of  Networks  ................  29 

15.  General  Principles  ....................  33 

NON-LINEAR  ELECTRIC  CIRCUITS 
BT  KNOX  MclLWAiN 

16.  Power  Series  Solution  .................  38 

17.  Trigonometric  Series  .................  39 


ART.  PAGE 

18.  Inductance  Variation 40 

19.  Capacitance  Variation 41 

20.  Approximate  Series  Expansion  for  the 

Plate  Current  of  a  Triode   (Assumes 

it  Constant) 41 

21.  Characteristics  of  Triode  with  Load 42 

22.  Analysis  for  Multi-electrode  Tubes 45 

23.  Method  of  Successive  Approximations . .  45 

24.  Harmonic  Analysis  of  the  Current  for  a 

Sinusoidal  Applied  Voltage 46 

25.  Input  Impedance  of  a  Triode 49 

ELECTROMAGNETIC  RADIATION 
BT  KNOX  MclLWAiN 

26.  Maxwell's  Equations 50 

27.  Progressive  Plane  Waves 51 

28.  Fields  Due  to  a  Current  in  a  Wire 52 

29.  Reflection  and  Refraction 52 

ELECTROMECHANICAL  SYSTEMS 
BT  KNOX  MclLWAiN 

30.  Energy    of    Mechanical    and    Electrical 

Systems 57 

31.  Vibrations  of  a  System  of  One  Degree  of 

Freedom 58 

32.  Comparison  of  Mechanical  and  Electrical 

Systems 59 

33.  Electromechanical-acoustic  Systems ....     62 


5-01 


ELECTRIC  CIRCUITS,  LINES,  AND  FIELDS 
THEORY  OF  LINEAR  PASSIVE  NETWORKS 

By  P.  H.  Richardson 

The  networks  to  be  considered  are  assumed  to  consist  of  resistances,  inductances, 
capacitances,  and  mutual  inductances  connected  or  coupled  together  in  some  manner. 
The  problem  is  to  determine  the  steady-state  response  of  the  network  to  an  impressed 
voltage,  or  current,  of  any  complexity.  It  is  presupposed  that  the  driving  voltage  has 
been  impressed  on  the  network  at  a  time  far  enough  in  the  past  to  have  permitted  any 
transients  to  die  out. 

To  facilitate  the  solution  of  the  problem  the  following  assumptions  are  made: 

1.  The  impressed  voltage  is  periodic. 

2.  The  network  is  linear.    The  values  of  the  component  elements  are  independent  of 
the  current  through  them. 

3.  The  network  is  passive.    There  are  no  sources  of  energy  interior  to  the  network,  and 
no  energy  is  dissipated  other  than  by  the  resistance  elements  of  the  network. 

1.  NON-SINUSOIDAL  CURRENTS  AND  VOLTAGES 

As  a  consequence  of  the  linearity  of  the  networks  the  coefficients  of  the  differential 
equations  are  real  constants.  The  equations  express  the  equilibrium  conditions  which 

exist  between  the  instantaneous  driving  voltages 

^  ^Jx^  "n  and  *ne  countervoltages  in  the  circuit.  For  ex- 

ample, in  the  circuit  of  Fig.  1,  consisting  of  R,  L, 
and  C  (=  1/D)  in  series,  the  differential  equation  is 


00 

FIG.  1.    Single-mesh  Circuit  where  q  is    the    charge    on   the    condenser,    and 

i  —  dq/dt.    If  <ji  is  the  charge  corresponding  to  an 

impressed  voltage  ei,  and  32  is  the  charge  corresponding  to  an  impressed  voltage  eg,  it  is 
evident  that 


and 

d^qz  dQz 

L  ~~~~  "~f~  R  ~ —  ~}~  DQ%  ==  ^2 

By  simple  addition  it  follows  that 

L  ^(Qldp  g2)  +  R  ^ti  ^  +  Dfa  +  &>  -  e>  +  e*  <2> 

Thus,  for  a  linear  network  the  principle  of  superposition  holds.  In  other  words,  the 
current  (or  voltage)  at  any  point  flowing  in  response  to  several  driving  voltages  acting 
together  is  the  sum  of  the  currents  (or  voltages)  at  that  point  which  would  flow  in  response 
to  the  driving  voltages  acting  separately.  This  principle  is  of  major  importance  in  network 
analysis. 

FOURIER'S  THEOREM.  A  second  important  concept  in  the  analysis  of  the  steady 
state  is  Fourier's  theorem,  which  states  that  any  single-valued  continuous  periodic  func- 
tion can  be  expressed  as  an  infinite  series  of  sine  waves.  In  particular,  if  f(x)  is  a  function 
which  is  finite  in  the  interval  from  —  c  to  +c  and  has  only  a  finite  number  of  discontinuities 
in  that  interval,  then  for  any  value  of  s  in  the  interval 

f,    ,  OQ      .  TTX     ,  27TX      , 

f(x)  =  —  +  ai  cos h  02  cos h  •  •  • 

,     -        .      7TX     ,     ,        .       2TTX     ,  ,„ 

+  6:801 1-62  sin h  •  •  •  (3) 

c  c 

5-02 


SINGLE-MESH  CIRCUIT  5-03 

where  the  coefficients  oo,  ai,  as,  •  •  •  61,  62,  •  -  •  are  determined  as  follows: 

1    rc  „  .         mrx  1     re  .    TITHE  _ 

an  =  ~  /    /(»)  cos  -  efo;         6W  =  -   I     /(x)  sin  -  da; 

CJ—C  C  C  J  —c  C 

Or,  equivalently, 

A*)  -  f  +  ^isin  (y  +  ft)  +  A2sHi  p?p  +  ft)  +  .••  (4) 

where 


A  proof  of  this  theorem  will  be  found  in  any  standard  book  on  calculus. 

This  theorem,  taken  in  conjunction  with  the  principle  of  superposition,  makes  it  pos- 
sible to  obtain  the  response  of  a  linear  network  to  a  periodic  voltage  of  any  complexity, 
provided  that  a  solution  is  available  for  the  case  in  which  the  driving  voltage  is  a  simple 
sinusoid.  The  complicated  wave  form  of  the  impressed  voltage  is  first  resolved  into  its 
component  sine  waves,  the  solution  of  the  equation  is  obtained  for  each  component,  and 
the  solutions  are  added  to  obtain  the  final  current  or  voltage  required. 

2.  SINGLE-MESH  CIRCUIT 

Before  proceeding  to  the  general  problem  it  will  be  profitable  to  examine  in  detail  the 
response  of  the  single-mesh  circuit  of  Fig.  1.  As  already  noted  the  differential  equation  is 


(5) 

air  at 

where  the  complex  exponential  form  has  been  written  for  the  impressed  voltage.  The 
notation  is  the  usual  one,  where  a?  ==  2-jrf,  f  —  frequency  in  cycles  per  second,  EQ  is  a 
constant,  either  real  or  complex,  and  y  —  V—  1. 

Assuming  a  solution  of  the  form  q  =  qoept  leads  to  the  result  that 

(Lp2  +  Rp  +  D^qoeP*  =  E«0»*  (6) 

which  is  evidently  a  solution  provided  that  p  —  ju.    Then 


go : 


and 


+  Rp  +  D 


where  IQ  ~  Eo/Z,  Z  =  R  +  Lp  +  D/p,  and  p  =  ja. 

Upon  substituting  ju>  for  p  the  result  is  the  familiar  one  that 

*'-|-V««  (8) 

Zi 

where  Z  —  R  4-  jx  and  x  —  (Leo  —  !>/&)  - 

The  use  of  the  complex  exponential  form  for  the  impressed  voltage  converts  the 
differential  equation  at  once  to  an  algebraic  equation.  The  resultant  current  is  a  complex 
exponential  form  of  the  same  frequency  as  the  impressed  voltage.  The  impressed  voltage 
is  made  up  of  two  components  in  quadrature,  and  the  resultant  current  is  also  made  up 
of  two  currents  in  quadrature.  Since  the  coefficients  of  eq.  (5)  are  real,  it  is  evident  that 
the  response  to  a  real  voltage  is  also  real,  while  the  current  in  response  to  an  imaginary 
voltage  is  imaginary.  Consequently,  by  the  principle  of  superposition,  the  physical 
current  flowing  in  response  to  a  voltage  EQ  cos  co£  is  given  by  the  real  part  of  eq.  (8)  . 
The  imaginary  part  of  the  current,  with  the  /  discarded,  is  the  physical  current  that 
flows  in  response  to  an  impressed  voltage  EQ  sin  ut.  Hence 


and 

%nag  -  T-T  sin  (tat  -f  a  —  j8) 


5-04 


ELECTRIC  CIRCUITS,  LINES,  AND  FIELDS 


where  Z  =  |  Z  \  /g,  EQ  =  |  EQ  \  /a,  and  0  «  tan"1  (a;/B) .  The  quantity  |  EQ  \  is  the 
maximum  value  of  et  and  similarly  |  Io  |  is  the  maximum  value  of  i.  By  proper  choice 
of  units  |  EQ  |  may  be  the  rms  voltage,  and  then  |  Jo  |  is  the  rms  current.  Hence  the  ratio 
|  EQ/IQ  I  becomes  the  absolute  value  of  the  steady-state  a-c  impedance.  The  complex 
quantity  Z  —  e/i  —  EQ/!Q  is  denned  as  the  complex  steady-state  impedance  and  may 
be  studied  as  a  function  of  p  =  jco. 


3.  THE  COMPLEX  FREQUENCY  PLANE 

The  definition  of  the  parameter  p  =  j2irf  can  be  usefully  extended  for  analytic  purposes 
to  situations  in  which  /  and  p  are  complex.     Suppose  a  voltage  E^cF*  is  assumed  where 
o  |  /«  and  p  =  pi  +  jpz.    Then 

Eo^  =  |  EQ  |  e*)iie7'^2*+a>  '    (10) 


In  accordance  with  the  above  discussion  the  physical  voltage  is  taken  as  the  real  part  of 
this,  which  is  a  sinusoidal  oscillation  with  positive  or  negative  damping  depending  on  pi. 
The  corresponding  steady-state  physical  current  is  obtained  by  dividing  the  complex 
voltage  by  the  impedance  and  taking  the  real  part  of  the  result.  This  is  a  damped  sinusoid 
of  the  same  frequency  and  damping  as  the  voltage. 

Complex  frequencies  can  be  represented  on  a  plane  as  shown  in  Fig.  2.    The  horizontal 
axis  represents  real  values  of  p,  and  the  vertical  axis  imaginary  values  of  p,  or  real  values 


1  8 


-Real  p  Axis 


p  plane 


+  Real  p  Axis 


If 

,33 


/R/  Y 

position  of 

v2Ly 

PI 

3?2 

0 

A 

A' 

+CO 

1 

<D/L 

B 

B' 

p  plane        _^J 

=  D/L 

G 

C 

B  ,' 

>D/L 

D 

D' 

,° 

/ 

^  1 

+  P 

D' 

~°\         D 

\ 

FIG.  2.    Complex  Frequency  Plane 


FIG.  3.    Distribution  of  Zeros  and  Poles 
in  Complex  Plane 


of  frequency.  In  network  analysis  a  distinction  of  primary  importance  is  made  between 
the  right  and  left  halves  of  the  p  plane,  since  on  the  left  half-plane  the  voltages  and  currents 
correspond  to  functions  which  decrease  exponentially  with  time,  whereas  on  the  right  half- 
plane  they  increase  exponentially  with  time.  There  is  a  close  connection  between  the 
steady-state  characteristics  of  a  network  and  its  transient  response.  Since  a  network 
whose  transients  increase  with  time  is  unstable,  the  characteristics  of  physical  networks 
in  the  right  half-plane  are  necessarily  limited.  There  is  no  such  distinction  between  the 
upper  and  lower  halves  of  the  plane. 

As  an  illustration  of  this  discussion  consider  the  impedance  of  the  single-mesh  network 
of  Fig.  1,    Here  the  impedance 

T   /  r>  n\ 

(ID 


where 


-  (P  ~  Pi)  (P  -  22) 


R 


are  the  zeros  of  the  impedance  expression.  They  are  seen  to  be  real  when  (B/2L)2  ^  D/L, 
while  they  are  conjugate  complex  numbers  when  (J2/2I/)2  <  D/L.  The  locations  of  the 
zeros  pi  and  pz  are  indicated  on  Fig.  3  as  R/L  is  varied  and  D/L  is  held  fixed.  It  should 
be  noted  that  the  values  of  pi  and  p%  always  have  negative  real  parts;  they  are  always  in 
the  left  half-plane.  The  impedance  becomes  infinite,  that  is,  it  has  a  pole,  when  p  is  zero, 


MESH  EQUATIONS 


5-05 


as  represented  by  the  cross  at  the  origin.    A  second  pole  occurs  at  infinity  where  the  im- 
pedance again  becomes  infinite. 


4.  MESH  EQUATIONS 

The  response  of  a  complicated  network  of  the  type  shown  in  Fig.  4  is  determined  by 
making  use  of  the  equilibrium  conditions  which  must  be  satisfied  by  the  instantaneous 
currents  and  voltages.  There  are  several  methods  of  writing  the  equations;  one  uses 
branch  equations,  a  second  mesh  equations,  and  a  third  nodal  equations.  In  writing 
branch  equations  the  current  in  each  branch,  ZIQ,  Z&,  etc.,  is  separately  specified,  and  the 
sum  of  the  instantaneous  voltages  in 
each  branch  is  equated  to  the  volt- 
age applied  to  the  ends  of  the  branch. 
There  are  B  such  equations,  where  B  is 
the  number  of  branches.  At  each  node, 
or  junction  point  between  branches, 
the  sum  of  the  currents  entering  the 
node  must  be  equal  to  the  sum  of  the 
currents  leaving  the  node.  Therefore, 
J  relations  of  this  kind  can  be  found, 
where  /  is  the  number  of  junction 
points.  If  there  are  S  separate  parts 
(not  conductively  connected  to  one 
another,  as  mesh  7  in  Fig.  4)  only 
J  —  S  of  these  equations  are  useful, 
since,  if  the  law  of  the  conservation 
of  charge  is  satisfied  at  all  but  one  node 
in  each  of  the  S  parts,  it  is  automat- 
ically satisfied  at  the  last  one  also. 
There  are  then  B  4-  J  —  S  independ- 
ent equations. 

The  original  branch  equations  in- 
clude, in  addition  to  the  B  branch  currents,  only  differences  in  the  node  voltages.  S  of 
these  can  be  arbitrarily  assumed,  and  there  will  remain  exactly  B  -f-  J  —  S  unknowns 
to  be  determined. 

By  adding  together  the  branch  voltage  equations  around  a  complete  loop,  or  mesh,  and 
eliminating  the  superfluous  branch  currents  by  means  of  the  nodal  current  equations,  an 
equation  is  found  for  each  mesh.  A  similar  equation  can  be  found  for  each  of  the  N 
meshes  of  the  network,  where  N  =  B  —  J  +  S. 

A  considerable  simplification  can  be  achieved  by  originally  specifying  only  one  current 
for  each  mesh  of  the  network  as  shown  in  Fig.  4.  By  choosing  circulating  currents  the 
nodal  equations  are  automatically  eliminated,  as  are  the  voltage  differences  which  appear 
in  the  branch  equations.  The  minimum  number  of  mesh  currents  that  can  be  used  is, 
of  course,  B  —  J  +  S.  For  example,  in  Fig.  4  there  are  15  branches,  10  junctions,  and  2 
parts,  hence  7  meshes.  It  is  important  to  note  that  a  closed  loop,  consisting  of  a  single 
branch  with  its  two  terminals  coinciding,  is  considered  to  have  one  junction. 

By  adopting  a  set  of  conventions,  regularity  of  notation  is  introduced  into  the  equations. 
Those  generally  used  are  that  all  currents  are  assumed  clockwise;  branches  common  to 
two  meshes  a  and  b  will  carry  as  subscript  the  symbols  of  both  meshes,  as  Za&;  branches 
appearing  in  one  mesh  only  will  be  designated  Za^\  and  the  sum  of  all  the  branches  in  a 
particular  mesh  will  be  designated  as  Zaa-  In  computing  Zaai  the  self-impedance  of  a 
mesh,  all  the  self-inductance  in  the  mesh  is  included  but  mutual  inductances  to  other 
meshes  are  not. 

Then  for  any  linear  network  of  N  meshes  a  set  of  N  equations  is  found  as  follows: 


2  70 
FIG.  4.    Multi-mesh  Network 


Zu.Ii  +  ^12/2  + 

Zzil*  +  #22/2  +  #23/3 


'  Zinln 


(12) 


where  each  of  the  Z's  is  of  the  form 

Zab  = 


and  p  and  1/p  represent  differentiation  and  integration  with  respect  to  time. 


5-06 


ELECTRIC   CIRCUITS,   LINES,   AND  FIELDS 


The  mesh  equations  developed  thus  far  have  represented  the  differential  equations  of 
the  circuit.  The  set  of  differential  equations  is  transformed  to  an  identical  set  of  algebraic 
equations  by  the  assumption  made  in  article  2,  that  each  of  the  sinusoidal  voltages  and 
currents  can  be  written  as  E^  or  ldwt,  where  E  and  I  are  constants,  co  =  2irf,  f  =  fre- 
quency in  cycles  per  second,  and  j  =  V  —  1.  This  transformation  results  from  the  fact 
that  differentiation  and  integration  of  e>ut  replaces  each  p  in  eq.  (12)  by  ju.  The  time 
factors  6»'w*  appear  on  both  sides  of  the  equation  and  can  be  divided  out.  The  Z's  are 
then  the  complex  self-  and  mutual  impedances  of  the  meshes.  And  the  7's  and  28*  &  are 
regarded  as  representing  merely  the  constant  coefficients  in  the  more  general  expressions 
JVW*  and  E^. 

DRIVING  POINT  AND  TRANSFER  IMPEDANCES.  The  determination  of  any 
particular  current  flowing  in  response  to  a  particular  voltage  is  equivalent  to  the  solution 
of  a  set  of  ordinary  linear  equations.  The  current  in  the  first  mesh  flowing  in  response  to 
the  voltage  Ei^wt,  also  in  that  mesh,  is  given  by  the  method  of  determinants  (see  Handbook 
of  Engineering  Fundamentals,  Eshbach,  John  Wiley)  as 


where  A  is  the  determinant  of  the  coefficients  in  the  left-hand  side  of  eq.  (12)  and  AH 
is  the  determinant  obtained  by  omitting  the  first  row  and  first  column  of  A.  The  driving 
point  impedance  in  the  first  mesh  is  by  definition  the  ratio  of  the  voltage  to  the  current 
in  eq.  (13).  Thus 


In  similar  fashion  the  current  in  the  second  mesh  flowing  in  response  to  the  voltage 
in  the  first  mesh  is  given  by 


(15) 


where  Ai2  is  the  minor  of  A  obtained  by  removing  the  first  row  and  second  column.    The 
transfer  impedance  from  the  first  to  the  second  mesh  is  defined  as 


(16) 


It  should  be  noted  also,  since  in  a  passive  network  Zab  =  Z&a,  that 

,    *     _*         A 


6.  NODAL  EQUATIONS 

An  analogous  system  of  equations  can  be  set  up  in  terms  of  driving  currents  at  the 

nodes  and  nodal  voltages.  In  this  analysis  the 
fundamental  equations  are  conditions  of  current 
equilibrium.  In  the  circuit  of  Fig.  5,  Ii,  /2,  /a,  and 
/4  are  the  driving  currents,  and  FH,  Fi2,  FIS,  etc., 
are  the  admittances  (reciprocal  impedances)  of  the 
various  branches.  Node  4  has  been  assumed  at 
ground  potential.  At  node  1,  then, 


(18) 


where  FU  =  FW  •+•  Fi2  +  FIS  is  evidently  the  sum 
of  the  admittances  of  all  the  branches  connected 
to  l  ™ih  aU  the  other  nodes  connected  together. 
FU  is  therefore  a  self-admittance  analogous  to  the 
self-impedance  in  the  mesh  analysis.  Similarly 


FIG.  5.    Illustration  for  Method  of  Nodal 
Analysis 


and  FIS  are  mutual  admittances  analogous  to  mutual  impedances.* 


*  For  a  treatment  of  nodal  analysis  for  circuits  involving  mutual  inductance  the  reader  is  referred 
to  Gardner  and  Barnes,  Transients  in  Linear  Systems,  VoL  I. 


TWO-TEEMINAL  IMPEDANCES  5-07 

In  any  conductively  united  network  having  /  nodes  a  set  of  J  —  1  independent  equa- 
tions of  the  above  form  can  be  written.    The  complete  set  becomes 

Ii 

J2  (19) 


+   YnnEn  -  In 

where  the  F's  are  of  the  form 

Yab  -  Cabp  +  0B6  +  -L  .  I 

Lab       P 

and  p  and  1/p  have  the  meanings  previously  ascribed  to  them. 

A  solution  of  the  set  of  nodal  equations  to  find  the  steady-state  voltage  corresponding 
to  a  given  set  of  sinusoidal  driving  currents  can  be  obtained  by  the  processes  already  used. 
The  driving  point  admittance  Y  between  the  first  node  and  ground  is  defined  as 


where  the  primes  indicate  that  the  A's  refer  to  the  set  of  equations  (19). 

Similarly  the  transfer  admittance  between  the  first  and  second  nodes  is  defined  as 

»-$--£ 

In  this  case,  also,  since  Yab  —  Yba, 

**-*--£--£ 

One  consequence  of  the  analogy  between  the  mesh  and  nodal  equations  is  that  the 
selection  of  one  of  the  two  methods  of  solution  in  any  particular  problem  is  entirely  a 
matter  of  convenience.  The  symmetry  of  the  two  methods  is  further  emphasized  by  the 
equivalence  of  Fig.  6,  in  which  a  constant-voltage  generator  in  series  with  an  impedance  Z 
is  shown  as  replaceable  by  a  constant-current  genera- 
tor in  parallel  with  an  impedance  Z. 

A  second  consequence  of  the  analogy  leads  to  the 
principle  of  duality  in  network  theory.  The  symmetry 
in  the  current  and  voltage  methods  of  analysis  in- 
cludes the  individual  terms  in  the  equations.  The 
general  term  Zab  of  eq.  (12)  is  replaced  by  Yab  of  eq. 
(19)  if 


Lab  —    Cab,  Rab  ss  Gab       and      Dab  —    (  73 —   I    —    ^ — 

\Cab/          Lab 

Consequently  a  set  of  nodal  equations  can  be  obtained 
identical  with  a  given  set  of  mesh  equations  by  inter- 
changing r  and  g,  and  L  and  C,  wherever  they  appear.  For  every  impedance  function, 
therefore,  there  is  a  corresponding  admittance  function.  If  the  mesh  equations  for  one 
network  correspond,  term  by  term,  with  the  nodal  equations  for  another,  the  two  networks 
are  called  inverse  structures,  or  duals. 

6.  TWO-TERMINAL  IMPEDANCES 

The  driving  point  impedance,  or  admittance,  of  a  network  can  be  expressed  as  the  ratio 
of  determinants  whose  elements  are  relatively  simple  functions  of  p  =  ju>.  In  the  mesh 
system  the  general  impedance  coefficient  can  be  written  as  Z0&  —  LabP  +  -Ka&  +  Dab  1/3?- 
Since  the  terms  A,  An,  and  Aaa  used  in  defining  driving  point  impedances  can  be  ex- 
pressed as  products  of  terms  of  this  type,  it  follows  that  they  are  polynomials  in  p  dividad 
by  some  power  of  p.  That  is, 


Bnpn  +  Bn-ip"-1  +  •  -  •  +  B1P  +  Bo 
Put  in  terms  of  the  zeros  and  poles  the  expression  is 

_        Am(p  -  pi)(p  -  #2)  •  •  •  (p  -  pm) 


(24) 


5-08  ELECTRIC   CIRCUITS,  LINES,   AND  FIELDS 

where  ordinarily  the  pa's  are  all  different.  Note  that  the  pa's  and  pa"s,  are  the  roots  of  the 
polynomials  in  the  numerator  and  the  denominator.  In  special  cases  two  or  more  zeros 
or  poles  may  coincide.  The  zeros  and  poles  may  be  thought  of  as  corresponding  to  the 
resonances  and  antiresonances  of  purely  reactive  networks  except  that  they  may  occur  at 
complex  frequencies. 

RESTRICTIONS  FOR  PHYSICAL  REALIZABILITY.  A  passive  two-terminal  or 
driving  point  impedance  is  subject  to  the  following  restrictions: 

1.  In  terms  of  the  frequency  variable  p  —  jco  the  zeros  and  poles  are  either  real  or  they 
occur  in  conjugate  complex  pairs. 

2.  The  real  and  imaginary  components  are  respectively  even  and  odd  functions  of 
frequency  on  the  real  frequency  axis. 

3.  None  of  the  zeros  and  poles  can  be  found  in  the  right  half  of  the  p  plane. 

4.  Zeros  and  poles  on  the  real  frequency  axis  must  be  simple. 

5.  The  real  component  of  the  driving  point  impedance  cannot  be  negative  at  real  fre- 
quencies. 

General  methods  for  finding  physical  networks  corresponding  to  any  impedance  func- 
tion meeting  these  restrictions  have  been  devised.*  A  method  due  to  Brune  depends  on 
the  fact  that  the  minimum  value  of  resistance,  or  conductance,  at  real  frequencies  is  less 
than  any  value  in  the  right  half-plane.  If  the  function  is  diminished  by  a  real  positive 
constant  equal  to  the  minimum  value  of  the  resistance,  the  remainder  corresponds  to  a 
passive  impedance  having  a  zero  resistance  at  a  real  frequency.  This  remainder  is  termed 
a  minimum  resistance,  or  minimum  conductance  expression.  Similarly  it  can  be  shown 
that  an  impedance  expression  having  zeros  or  poles  on  the  real  frequency  axis  can  be 
diminished  by  the  reactance,  or  susceptance,  corresponding  to  its  real  frequency  zeros  and 
poles.  An  impedance  having  no  poles  at  real  frequencies  is  called  a  minimum,  reactance 
expression,  while  one  having  no  zeros  at  real  frequencies  is  called  a  minimum,  susceptance 
expression.  If  an  impedance  is  both  minimum  resistance  and  minimum  reactance,  there 
is  a  unique  relation  between  the  resistance  and  reactance;  if  either  is  known  at  all  fre- 
quencies the  other  can  be  determined. 

NETWORKS  OF  PURE  REACTANCES.  If  it  is  specified  that  the  zeros  and  poles  of 
an  impedance  expression  occur  at  real  frequencies,  or  imaginary  values  of  p,  the  form 
of  the  impedance  expression  becomes 

~   P22)(P2  ~   P42)    •  '  '    (P2  ~   Pm2)  ,       . 


= 

(P2  ~   Pi2)  (P*  ~   P32)    '  •  '    (P*  ~   Pm-fi 

where  k  is  a  positive  real  constant,  while  pi2,  p22,  etc.,  are  negative  real  quantities.  Each 
of  the  factors  represents  a  pair  of  zeros,  or  poles,  at  positive  and  negative  real  frequencies. 
The  zeros  and  poles  are  restricted  in  that 

-  Pm2  ^    ~  Pm-l2  ^    ~    '  -  '    ^    ~   P22  ^    ~   Pi2  ^   0  (26) 

or  the  zeros  and  poles  must  alternate. 

As  written  the  impedance  is  specified  as  an  inductive  reactance  at  both  zero  and  infinite 
values  of  frequency.  To  obtain  complete  generality  it  must  be  allowable  to  specify 
p^  =  o  (which  introduces  a  pole  at  p  =  0),  or  that  the  factor  p2  —  pm2  can  be  omitted 
(which  leads  to  a  zero  at  infinite  frequency)  . 

It  has  been  demonstrated  f  that  the  impedance  of  eq.  (25)  corresponds  to  a  physical 
network  containing  only  inductances  and  capacitances.  The  network  can  be  found  either 

representing  the  poles  as  antiresonant  net- 


I-MTOQT>     |        -     qnnp     1       works  in  series  or  by  representing  the  zeros  as 
JL   _L  _J_,    resonant  networks  in  parallel.    For  a  detailed 

~T  "T   ~T  ~T"    discussion  of  these  impedances  see  Section  6, 


,  s  -^-  _u_   _j_  __i_    resonant  networks  in  parallel.    For  a  detailed 

(a)  T  T    T  T< 

Article  17. 

The  reactive  networks  can  also  be  realized 
in  other  configurations  of  which  Figs.  7a  and 
76  are  typical.  These  are  obtained  as  the  re- 
sult of  continued  fraction  expansions  in  which 
the  zeros  and  poles  at  zero  and  infinite  frequency 
are  removed  alternately. 


FIG.  7.    Ladder-type  Reactive  Networks  c^e  o    the  restrictions  on  the  zeros  and 

poles  the  impedance  characteristics  of  reactive 

networks  are  necessarily  restricted.  The  slope  of  the  characteristic  is  always  positive  and 
necessarily  greater  than  that  of  a  simple  inductance  or  capacity  having  the  same  reactance 
at  a  given  frequency. 

*  Brune,  Journal  of  Mathematics  and  Physics,  M.I.T.,  Vol.  X,  October  1931,  pp.  191-235.    Darling- 
ton, Journal  of  Mathematics  and  Physics,  M.I.T.,  Vol.  XVIII,  No.  4,  September  1939,  pp.  257-353. 
tR.  M.  Foster,  A  Reactance  Theorem,  B.S.T.J.,  April  1924,  pp.  259-267. 


TWO-TERMINAL  IMPEDANCES 


5-09 


NETWORKS  OF  RESISTANCES  AND  INDUCTANCES  OR  RESISTANCES  AND 
CAPACITANCES.  Very  similar  to  the  purely  reactive  networks  wliich  result  when  the 
zeros  or  poles  are  specified  at  real  frequencies,  a  simple  series  of  networks  results  when  it 
is  specified  that  the  zeros  and  poles  occur  at  imaginary  frequencies,  or  real  values  of  p. 
Again  only  two  kinds  of  elements,  either  resistances  and  inductances  or  resistances  and 
capacitances,  are  required  to  realize  such  impedances. 

If  the  expression  corresponds  to  a  network  of  resistances  and  inductances,  it  is  of  the  form 


kp(p  — 


•  •  >  (p  — 


•  •  •  (p  - 


(27) 


and  |  Z  |  increases  as  p  increases.  If,  on  the  other  hand,  the  expression  corresponds  to  a 
network  of  resistances  and  capacitances  |  Z  \  decreases  as  p  increases.  An  alternative 
form  for  Z  is 

k(p  —  02)  (p  —  ai)  -  •  •  (p  —  am)  . 

~  -  ~  -  -  - 


~, 

—   ai)(p  — 


—   Om-l) 


In  both  expressions  the  zeros  and  poles  occur  alternately  and  ai,  02,  etc.,  are  negative  real 
quantities  or  zero. 

As  in  the  case  of  the  reactive  network  the  impedances  may  be  represented  in  partial 
fraction  form.    For  example,  the  impedance  of  eq.  (28)  can  be  expanded  in  the  form 


Z  « 


P  — 


, 
•  -r  " 


p  — 


Dm-l 


(29) 


where  Da  =  [(p  —  aa)Z]p=aa  and  R  is  the  resistance  at  infinite  frequency.  Each  of  the 
terms  Da/(p  —  aa)  is  identifiable  with  the  parallel  combination  of  resistance  and  capaci- 
tance, where  ra  —  —  Da/aa  and  Ca  =  I/At- 

INVERSE  OR  RECIPROCAL  IMPEDANCES.  The  duality  between  the  impedance 
and  admittance  methods  of  analyzing  a  network  suggests  the  possibility  that  to  every 
network  there  corresponds  an  inverse.  The  requirement  that  the  real  part  of  an  im- 
pedance be  positive  is  merely  another  way  of  stating  that  the  real  part  of  the  corresponding 
admittance  be  positive.  Also,  the  restrictions  on  the  zeros  and  poles  are  identical,  so  that 
the  interchange  of  zeros  and  poles  when  an  impedance  is  replaced  by  its  reciprocal  does 
not  change  the  conditions  for  physical  readability.  It  follows,  then,  that,  if  a  passive 
impedance  is  physically  realizable,  its  reciprocal  is  also. 
^  The  reciprocal  impedance  for  the  structure  of  Pig.  8a  is  found,  for  example,  as  follows: 

1.  Each  series  connection  is  replaced  by  a  parallel  connection,  and  vice  versa. 

2.  The  individual  resistances,  inductances,  and  capacitances  are  respectively  replaced 
by  resistances,  capacitances,  and  inductances  in  such  a  way  that 


The  structural  inverse  of  Fig.  8a  is  therefore  given  by  86.    This  process  is  evidently  not 
general  since  it  considers  only  series  and  parallel  connections.    An  extension  of  the  method 


FIG.  8.    Inverse  Two-terminal  Networks 


FIG.  9.    Inverse  Bridge  Networks 


depends  on  a  consideration  of  the  geometry  of  the  network.    The  branches  of  the  network 
are  considered  as  lines  between  the  junction  points,  dividing  the  plane  of  the  diagram  into 


5-10 


ELECTRIC   CIRCUITS,   LINES,   AND  FIELDS 


areas.     The  process  consists  of  interchanging  areas  and  points.     A  new  point  is  taken 
interior  to  each  area  and  is  connected  to  each  similar  point  by  a  branch  inverse  to  the 
branch  separating  the  areas.     The  inverse  of  a  bridge  network  is  found  by  this  process 
to  be  another  bridge,  as  shown  in  Fig.  9.    Even  this  process  is  not  entirely  general.* 
COMPLEMENTARY  IMPEDANCES.     In  addition  to  the  inverse  of  a  given  impedance 

function  one  can  also  speak  of  its  com- 
plement. The  complement  is  denned  by 
the  requirement  that  the  sum  of  the 
original  impedance  and  its  complement 
is  a  real  constant.  The  complement  of 
a  passive  impedance  can  be  found  if  the 
prescribed  impedance  has  no  poles  on 
the  real  frequency  axis,  and  if  the  sum 
of  the  impedance  and  its  complement  is 
chosen  at  least  as  great  as  the  maximum 
value  of  the  resistance  of  the  original  impedance.  The  constant  resistance  combination  of 
Fig.  10  represents  a  simple  example  of  the  relationship. 


when  —  =- 
FIG.  10.    Complementary  Impedances  in  Series 


7.  FOUR-TERMINAL  NETWORKS 

The  four-terminal  network,  or  two-terminal  pair,  is  a  special  form  of  general  network 
of  major  importance.    The  external  characteristics  of  the  network  are  completely  specified 
in  terms  of  Ii,  EI,  /2,  and  E%  of  Fig.  11.    A  solution  is  ob- 
tained from  eq.  (12)  with  the  assumption  that  all  voltages         ll  -  2 
except  EI  and  E%  are  zero.    Thus                                                1  <; 


An 
— 


A22 

__ 


(30a) 


(306)     Fia.  11.     General  Four-terminal 
Network 


Or,  solving  explicitly  for  EI  and  E%,  and  noting  that  AAn22  =  AnA22  —  Aig2 

EI  =  —  /i  +  -^- 12  (31a) 

Ez  =  —  I  +  An  I2  (316) 

The  currents  of  eqs.  (30)  are  determined  by  the  voltages  EI  and  E%  and  the  quantities 
An/A,  A22/A,  and  —  Ai2/A. 

DRIVING  POINT  AND  TRANSFER  IMPEDANCES.     The  physical  significance  of 
the  ratios  is  seen  by  successively  setting  Ez  and  EI  equal  to  zero.    Thus 

-^  =  Ysi  =  admittance  at  terminals  1-1'  with  2-2'  shorted 


22 

—  =  Yaz  =  admittance  at  terminals  2-2'  with  1-1'  shorted 
— —  M 


YS2i  ~  transfer  admittance  from  either  end  with  the  opposite  end  shorted 
Similarly,  if  1$  and  Ji  of  eqs.  (31a)  and  (316)  are  successively  set  equal  to  zero, 
'  impedance  at  terminals  1—1'  with  2-2/  open 

An 


A22  „ 

- —  —  &oi 

An-22 


a  =  impedance  at  terminals  2-2'  with  1-1'  open 
-~-  =  Zoi2  =  ^021  ~  transfer  impedance  from  either  end  with  the  opposite  end  open 


Thus  the  network  may  be  described  by  either  of  these  sets  of  three  parameters,  or  by 
other  sets  of  three  parameters  properly  related  to  them. 

*  Foster,  Geometrical  Circuits  of  Electrical  Networks,  Trans.  A.I.E.E.,  June  1932. 


FOTJE-TEEMINAL  NETWORKS 


5-11 


The  driving  point  impedances  and  admittances  are  subject  to  the  same  restrictions 
as  any  other  two-terminal  networks,  if  they  are  to  correspond  to  physical  networks.  The 
transfer  functions,  however,  differ  in  several  impor- 
tant respects  and  require  further  consideration. 

For  the  terminated  network  of  Fig.  12  it  can  be 
shown  that 


(*+£)• 


(32o) 


(326)     FIG.     12.      Terminated    Four-terminal 
Network 

The  response  of  the  network  evidently  depends  on  the  terminations  Z\  and  Zo,  as  well  as 
on  the  network  parameters  themselves.    Thus  the  driving  point  impedances  become 


(33) 


(34) 


An  • 
And  the  transfer  impedances  ZT  =  Er/I\ 

ZT  =  -  ^~  [A  -f  2 


A22 


The  form  of  the  transfer  impedance  expression  is  seen  to  be  the  ratio  of  determinants 
(since  eq.  [34]  can  be  written  ZT  —  —  A/Ais  if  A  is  understood  to  be  the  network  determi- 
nant including  the  terminations  Z\  and  Z%),  The  general  impedance  coefficient  is 
Zab  —  LabP  +  Rab  +  Dab(l/p)  as  before.  In  terms  of  the  frequency  variable  p  —  /<o, 
therefore,  the  transfer  impedance  is  the  ratio  of  polynomials  in  p  and  can  be  written  in 
terms  of  its  zeros  and  poles  with  a  constant  multiplier,  just  as  in  the  case  of  driving  point 
impedances.  Thus 

Z  A*(P  ~    ^(P  -    OQ    '  '  '    (P   -    O  ,x 

T 


Bn(p  - 


-  62)  -  -  -  (p  -  bn) 


Since  ZT  represents  a  transmission  it  is  usually  stated  as  a  logarithm.  And  also,  since 
the  most  efficient  possible  transmission  between  two  impedances  Z\  and  Zi  is  obtained  if, 
first,  the  reactances  of  Z\  and  Z^.  are  annulled,  and  then  a  transformer  of  optimum  ratio  is 
inserted  between  the  two,  this  condition  is  used  as  a  reference.  Then  the  general  transfer 
impedance  is  _ 

ZT  =  2VRiR&8  (36) 

where 

--         •"        -  (37) 


(P  -  &i)(p-  62)  •  •  •  (p  -  W 

and  RI  and  RZ  are  the  real  parts  of  Z\  and  Z%.  The  o's  and  5's  are  the  zeros  and  poles  of  ZT, 
or  the  points  of  infinite  gain  or  loss  in  terms  of  9.  It  is  evident  from  eq.  (37)  that  two 
transfer  impedances  having  the  same  zeros  and  poles  can  differ  only  by  a  constant  loss 
or  gam. 

RESTRICTIONS  FOR  PHYSICAL  REALIZABILITY.  The  restrictions  which  must 
be  met  if  ZT  is  to  correspond  to  a  physical  network  are  as  follows: 

1.  In  terms  of  the  frequency  variable  p  both  the  zeros  and  the  poles  must  be  real  or 
must  occur  in  conjugate  complex  pairs. 

2.  The  real  and  imaginary  components  are  respectively  even  and  odd  functions  of  fre- 
quency. 

3.  The  zeros  must  be  located  in  the  left  half  of  the  p  plane;  the  poles  may  occur  in  any 
part  of  the  plane. 

4.  The  real  part  of  0  —  A  4*  jB  is  positive  at  real  frequencies;  otherwise,  the  network 
serves  as  a  source  of  power. 

On  comparing  these  restrictions  to  those  given  for  two-terminal  impedances  two  im- 
portant differences  are  noted.  First,  the  poles  of  the  transfer  impedance  are  not  restricted 
to  the  left  half-plane,  and  second,  the  real  part  of  the  transfer  impedance  may  be  negative. 
The  previous  restriction  that  the  real  part  of  a  driving  point  impedance  be  positive  is 
replaced  by  the  new  restriction  that  the  real  part  of  &  must  be  positive. 


5-12 


ELECTEIC   CIKCUITS,   LINES,   AND  FIELDS 


A  structure  which  may  be  used  to  represent  the  general  passive  transfer  function  is 
shown  in  Fig.  13.  The  arms  of  the  symmetrical  lattice  are  assumed  to  be  inverse  such 
that  Zy.Zy  =  R#.  For  this  structure  the  ratio 


where 


Zx 


Zx 


(38) 
(39a) 

(396) 


It  can  be  shown  that 


FIG.  13.    Network  Having  Any  Prescribed  Pas- 
sive Transfer  Function 


to  phase-shifting  networks, 
the  combinations  of  factors 

p  +  a 
p  -  a 


represents  a  physical  impedance  as  long  as  0  satisfies  the  re- 
strictions listed  above.  In  particular,  Zx  will 
be  physical  as  long  as  the  transfer  loss  A  is 
greater  than  zero  at  real  frequencies.  Con- 
sequently, in  any  case  hi  which  the  minimum 
value  of  A  is  finite,  the  loss  may  be  reduced 
by  a  constant  corresponding  to  this  minimum 
value.  The  reduced  expression  having  zero 
loss  at  a  real  frequency  is  called  a  minimum 
loss  or  minimum  attenuation  function. 

It  is  also  possible  to  modify  the  transfer 
phase  B  without  affecting  the  loss  by  intro- 
ducing, or  eliminating,  terms  corresponding 
Physical  networks  can  be  found  (article  10)  corresponding  to 


and 


(p  —  a  —  jb}  (p  —  a  +  jb) 


(40) 


Note  that  the  poles  in  these  expressions  occur  in  the  right  half  of  the  p  plane,  and  that 
the  zeros  and  poles  are  the  negatives  of  one  another.    The  absolute  value  of  these  expres- 
sions is  evidently  unity  for  all  values  of  p,  but  the  phase  angle  depends  on  p. 
If  the  function  ZT  contains  a  single  pole  on  the  positive  real  axis, 


ZT 


F(P) 
p  -  a       p  + 


F(p)    ip  +  a\ 
3  +  a  \p  -  a) 


(41) 


The  transfer  impedance  ZT  is  thus  the  product  of  a  new  transfer  impedance  ZT'  and  a 
phase-shifting  term,  and  can  be  shown  to  correspond  to  a  new  network  of  transfer  im- 
pedance ZT'  in  tandem  with  a  simple  phase  network.  A  similar  treatment  is  possible  for 
a  pair  of  complex  poles  in  the  right  half-plane.  The  modified  transfer  impedance  having 
no  poles  in  the  right  half-plane  is  termed  a  minimum  phase  function.  No  further  reduction 
can  be  made  in  the  phase  characteristic  of  such  a  function  without  at  the  same  time 
affecting  the  loss  characteristic. 

NETWORK  THEOREMS.  Several  useful  theorems  can  be  stated  for  the  general 
linear  network  as  follows: 

The  Compensation  Theorem.  If  an  impedance  AZ  is  inserted  in  a  branch  of  a  network 
the  resulting  current  increment  produced  at  any  point  in  the  network  is  equal  to  the  cur- 
rent that  would  be  produced  at  that  point  by  a  compensating  voltage  — 7A.Z  acting  in 
series  with  the  modified  branch,  where  /  is  the  current  in  the  original  branch. 

The  Reciprocity  Theorem.  If  an  electromotive  force  E  of  zero  internal  impedance 
applied  between  two  terminals  of  a  network  produces  a  current  I  in  some  branch  of  the 
network,  then  the  same  voltage  E  acting  in  series  with  the  second  branch  will  produce  the 
current  I  through  the  first  pair  of  terminals  shorted  together.  This  follows  from  eqs.  (30) 
since  the  short-circuit  transfer  admittance  is  the  same  from  either  end  of  the  network. 

Thevenin's  Theorem.  With  respect  to  any  pair  of  terminals  considered  as  output 
terminals  the  network  can  be  replaced  by  a  branch  having  an  impedance  Z&,  equal  to 
the  driving  point  impedance  at  these  terminals,  in  series  with  an  electromotive  force  E, 
equal  to  the  open-circuit  voltage  across  these  terminals.  An  analogous  theorem  can  be 
expressed  in  terms  of  the  short-circuit  current  entering  the  output  node  of  the  network. 


FOUR-TERMINAL  NETWORKS  5-13 

These  results  follow  from  eq.  (306)  and  the  definitions  of  the  open-circuit  impedances  and 
short-circuit  admittances.    Thus,  if  EZ  =  —  IzZz, 


(42) 


(43) 


Za)       ZQl(Z32  +  Za) 
The  open-circuit  voltage  at  terminals  2  —  2'  is  —  I«Zz  when  £2  —  »  °°  ,  whence 


The  short-circuit  current  at  terminals  2  —  2'  is  —  Ei  Y8iz,  whence 

l2' 


EQUIVALENT  QUADRIPOLES.  A  useful  concept  in  network  analysis  is  that  of 
"equivalence."  Two  four-terminal  networks,  or  quadripoles,  are  considered  to  be  equiva- 
lent when  the  fundamental  relations  describing  the  behavior  of  the  networks  with  respect 
to  their  input  and  output  terminals  are  identical.  Such  equivalences  can  be  expressed 
in  terms  of  the  network  determinant  and  its  minors,  the  open-circuit  impedances,  the 
short-circuit  admittances,  or  any  other  set  of  three  convenient  and  properly  related 
parameters. 

IMAGE  PARAMETERS.  An  important  and  useful  set  of  parameters  is  based  on 
the  idea  that  the  terminations  of  the  network  be  so  related  to  the  network  itself  that 
the  impedances  looking  in  both  directions  from  the  input  terminals,  or  in  both  directions 
from  the  output  terminals,  are  the  same.  Referring  to  Fig.  12  and  eq.  (33)  this  means 
that  Zi-i  =  Zi  and  £2-2'  =  Zz.  The  impedances  Zi  =  Z/:  and  Z2  =  Z/2  are  called 
image  impedances  and  are  functions  of  the  network  itself,  since 

and     Z/a  = 

The  third  parameter  necessary  to  characterize  the  network  is  called  the  image  transfer 
constant  and  is  defined  as 

That  is,  6  is  one-half  the  logarithm  of  the  ratio  of  the  volt-amperes  entering  the  network 
to  the  volt-amperes  leaving  the  network  when  it  is  terminated  in  its  image  impedances. 
In  terms  of  the  network  determinant 


The  value  of  these  parameters  lies  in  the  fact  that  they  offer  approximations  to  the  actual 
network  behavior  and  serve  to  relate  the  behavior  of  a  network  to  that  of  a  transmission 
line. 

T  AND  IT  NETWORKS.  The  interrelation  of  several  sets  of  network  parameters  is 
shown  for  two  fundamental  types  of  four-terminal  networks  in  Fig.  14.  The  branches  of 
the  T  and  TT  networks  equivalent  to  the  general  network  are  given  in  terms  of  the  network 
determinant  and  its  minors,  the  open-  and  short-circuit  impedances,  and  the  image 
parameters.  These  branch  impedances  and  admittances  themselves  constitute  complete 
sets  of  network  parameters,  even  though  they  may  not  be  physically  realizable  as  two- 
terminal  networks  at  all  frequencies. 

The  T  and  TT  networks  are  important  since  they  can  be  used  to  represent  any  quadripole 
for  purposes  of  analysis  and  computation.  They  may  evidently  be  used  to  represent  any 
three-terminal  network,  or  segment  of  a  network,  without  loss  of  generality.  The  L-type 
network  is  considered  to  be  a  degenerate  case  of  the  T  or  TT  networks,  and  it  is  significant 
in  that  only  two  independent  parameters  are  required  to  specify  its  behavior. 

LATTICE  OR  BRIDGE  NETWORKS.  A  network  of  major  importance  is  the  sym- 
metrical balanced  lattice  shown  in  Fig.  15.  It  can  be  shown  that  this  network  is  the 
most  general  of  all  symmetrical  networks,  and  that  any  passive  symmetrical  four-terminal 
network,  or  quadripole,  can  be  represented  by  a  physically  realizable  lattice.  Further, 
since  the  image  impedance  depends  only  on  the  product  and  the  image  transfer  constant 
only  on  the  ratio  of  the  branch  impedances,  it  is  possible  to  control  the  transmission  and 
impedance  characteristics  independently.  See  Section  6,  article  23. 


5-14  ELECTRIC   CIRCUITS,   LINES,   AND   FIELDS 


tanh  6  =» 


=  ZA  +  Zc  - 


YF 


Symmetrical:  (An  =  A22t  -Z"oi 


—  ZJB  =  Z7  tanh  - 


,  =  YE  =  YI  tanh  - 
Yl 


FIG.  14.    T  and  ir  Networks  Equivalent  to  a  General  Dissymmetrical  Network 

2Z, 


An  -  Aig 


tanh  5/2 


Zi  —  v'zxZy     tanh  -  =-   \  — 


FIG.  15.     Lattice  Network  Equivalent  to  Any  Symmetrical  Network  and  to  T  and  Bridged-T  Net- 
works 


POWER  TRANSFER  5-15 

The  lattice  equivalent  to  the  symmetrical  T  and  bridged  T  structures  of  Fig.  15  is  seen 
to  be  physical  as  long  as  the  branch  impedances  of  these  structures  are  physical.  The  con- 
verse is  not  necessarily  true,  since  each  of  these  networks  requires  that  the  arms  of  the 
lattice  contain  a  common  impedance. 

8.  POWER  TRANSFER 

Transmission  through  a  network  is  usually  expressed  as  a  logarithm  with  respect  to  a 
suitable  reference  condition. 

Transition  Loss.  The  condition  for  maximum  power  transfer  from  a  generator  to  a 
load  is  indicated  in  Fig.  16.  The  series  reactances  —  JS"i  and  —  Xz  are  inserted  to  annul 
the  corresponding  reactances  of  the  generator  and  load  impedances,  and  the  ideal  trans- 
former of  optimum  ratio  matches  the  resistance  of  the  generator  to  that  of  the  load. 
This  structure  is  termed  an  ideal  transducer,  and  the  loss  in  power  which  is  eliminated  when 
it  is  inserted  between  a  generator  and  a  load  is 
called  the  transition  loss  or  the  transducer  loss.  In 
decibels  the 

T> 

Transition  loss  —  10  logio  -~ 
Pz 

=  20  logio  |  Zi  +  Z2  |  -  10  logio  4R&     (47) 
where  Pso  =  EP/^Ri  is  the  reference  power  in  the 


load  ( =  available  power)  and  P2  =  ffRz/l  Zi  +  Z2 12  FIG.  16.    Ideal  Transducer 

is  the  actual  power  in  the  load. 

Insertion  Loss  and  Phase.  When  a  network  is  inserted  between  a  sending  impedance  Z\ 
and  a  receiving  impedance  Z^  a  change  occurs  in  the  current  (or  voltage)  in  the  load. 
The  ratio  of  the  original  load  current  120  to  the  new  load  current  /2  is  denned  as  the  insertion 
loss  factor  or  the  insertion  factor.  In  terms  of  the  image  parameters  this  becomes 

•*20  -p 

i;  =  <T 

where  , 


fa  =  -— — -1  =  the  sending  end  reflection  factor 

Z\  +  Zi^ 

2VZzZi2 


fa  = — -  =  the  receiving  end  reflection  factor 

fc  —  L_?  3—  the  reflection  factor  between  Z\  and  Z% 

0  =  the  image  transfer  constant 

S  =  ~ 7^ ^-TTTZ ^— ; T  =  the  interaction  factor 


The  reflection  factors  represent  modifications  in  the  load  current  caused  by  reflections 
at  the  input  and  output  junctions.  The  factor  k,  sometimes  called  the  symmetry  factor, 
represents  a  reflection  factor  that  was  eliminated  when  the  network  was  inserted.  Each 
of  these  factors  involved  only  the  ratio  of  two  impedances,  and  each  becomes  unity  when 
the  impedances  are  equal. 

The  interaction  factor,  S,  is  a  second-order  effect  which  takes  account  of  a  wave  reflected 
from  the  load  back  to  the  generator  and  then  back  to  the  load.  It  reduces  to  unity  when 
either  termination  matches  the  image  impedance  adjacent  to  it,  or  when  the  attenuation 
of  the  network  is  high. 

The  insertion  factor  becomes  equal  to  efl  when  one  of  the  terminating  impedances  and  the 
image  impedances  are  equal  to  one  another. 

The  insertion  loss  (in  decibels) 


=  20  logio 

-  2o|~  -  logio  i4r  +  logic  r^-r  +  logio  r^-r  +  logio  r^T  +  IQ&O  t  £  1 1      (49) 

L  I « I  l^il  i«2l  1^1  -I 


5-16  ELECTRIC   CIRCUITS;   LINES,   AND   FIELDS 

The  first  three  terms,  involving  the  reciprocals  of  the  reflection  factors,  are  called  reflection 
losses.  The  fourth  term  is  called  the  interaction  loss,  and  the  last  term  represents  the  real 
part  of  the  image  transfer  constant  in  decibels, 

The  insertion  phase  shift  is  the  phase  angle  of  the  current  ratio  given  by  eq.  (48). 

9.  DISTORTION 

When  a  voltage  of  complicated  wave  form  is  introduced  into  an  electric  circuit,  a  current 
will  flow  whose  wave  form  will  depend  on  that  of  the  voltage  and  on  the  transmission 
characteristics  of  the  circuit  itself.  It  is  frequently  desired  that  the  wave  form  of  the  cur- 
rent through  a  particular  circuit  element  shall  be  the  same  as  the  wave  form  of  the  original 
voltage.  If  the  complicated  impressed  voltage  is  regarded  as  a  series  of  sine  waves  of 
various  frequencies  the  conditions  under  which  the  output  current  will  be  a  faithful 
reproduction  of  the  input  voltage  may  be  stated  as  follows  : 

1.  The  response  of  the  circuit  must  be  the  same  for  all  frequency  components  present 
in  the  impressed  voltage. 

2.  The  relative  phase  relations  of  the  various  frequency  components  must  not  be 
altered. 

3.  The  circuit  must  be  linear. 

FREQUENCY  DISTORTION.  When  the  first  condition  is  not  satisfied  and  the  rela- 
tive amplitudes  of  the  various  frequency  components  are  altered,  it  is  said  that  frequency 
distortion  occurs.  This  type  of  distortion  occurs  when  the  voltage-current  characteristic, 
that  is  the  transfer  impedance  characteristic,  is  a  function  of  frequency.  Where  the  whole 
transmission  apparatus  (including  any  mechanical  and  acoustic  portions)  is  considered, 
there  are  two  methods  by  which  the  effect  may  be  minimized.  Each  component  part  of 
the  system  may  be  so  designed  as  to  have  its  response  independent  of  frequency,  or  some 
elements  of  the  system  may  be  designed  to  correct  for  the  distortion  introduced  elsewhere. 
Both  methods  of  design  have  been  widely  and  successfully  used,  the  choice  for  a  particular 
case  being  decided  usually  by  economic  considerations. 

DELAY  DISTORTION.  The  requirement  that  the  relative  phases  of  the  various 
components  be  unaltered  is  equivalent  to  the  requirement  that  the  time  of  transmission 
of  the  system  be  independent  of  frequency.  To  see  this,  consider  a  voltage  consisting  of 
a  group  of  sine  waves  applied  to  the  system.  In  complex  notation 


«  -  23  JBdW+to  (50) 

fcl 

The  received  current  will  be 


*i 

where  Zk  ~  \  Zk  \  f&k  is  the  transfer  impedance  of  the  circuit  at  each  frequency  &k/2ir. 
It  is  evident  that  frequency  distortion  will  be  present  if  |  Zk  \  is  a  function  of  frequency 
within  the  band  of  frequencies  considered.     If  it  be  assumed  that  [  Zk  \  =  R  and  that 
Pk  =  «$i  =fc  nir,  where  R  and  t\  are  constants  and  n  =  0,  1,  2,  •  •  •  ,  then 

,   to 

*  -  ±  4  S  &#*»*<*-**  +dk]  (52) 

R  *i 

The  effect  of  the  transfer  impedance  is  to  delay  each  component  by  an  amount  ti  but  to 
leave  the  relative  phases  unaltered.  The  change  in  sign  which  occurs  when  n  is  odd  is 
usually  not  important. 

If,  on  the  other  hand,  (3%  —  &kti  +  <r(u>)  ±  mr,  where  <r(&>)  is  not  a  linear  function  of 
frequency,  the  wave  form  of  the  received  current  will  be  different  from  that  of  the  impressed 
voltage  even  though  the  relative  amplitudes  of  the  current  components  are  correct.  (For 
a  discussion  of  non-linear  distortion  see  Non-linear  Electric  Circuits.) 

10.  CORRECTIVE  NETWORKS 

A  corrective  network,  or  equalizer,  is  a  network  inserted  between  a  generator  and  a 
load  such  that  the  current  in  the  load  will  vary  with  frequency  in  a  predetermined  manner. 
A  loss,  or  attenuation,  equalizer  is  one  which  is  used  to  control  the  amplitude  of  the  received 
current  as  a  function  of  frequency  without  regard  to  phase  relations.  A  phase  equalizer 


CORRECTIVE  NETWORKS 


5-17 


is  one  which  ideally  introduces  no  loss  but  does  introduce  phase  shift  as  a  function  of 
frequency. 

In  most  cases  it  is  sufficient  to  equalize  only  for  changes  in  loss.  If,  however,  both 
loss  and  phase  equalization  are  required,  it  is  usual  first  to  equalize  for  loss,  and  then  to 
correct  the  phase  of  the  system  plus  the  loss  equalizers.  The  incidental  loss  characteristic 
introduced  by  the  phase  equalizer  (because  of  power  dissipated  in  the  ideally  reactive 
elements)  is  usually  ignored.  If  necessary,  this  distortion  is  corrected  by  an  additional 
loss  equalizer  designed  as  an  integral  part  of  the  phase  equalizer. 

LOSS-PHASE  RELATION.  It  is  generally  true  that  no  unique  relation  can  exist 
between  the  loss  and  phase  characteristics  of  a  four-terminal  network.  However,  as  noted 
in  article  7,  there  is  a  unique  relation  between  a  given  loss  characteristic  and  the  minimum 
phase  shift  that  can  be  associated  with  it.  This  relation  is  of  value  in  the  design  of  correc- 
tive networks  and  feedback  amplifiers,  where  it  is  necessary  to  control  both  loss  and  phase 
over  wide  frequency  ranges. 

For  the  minimum  phase  condition  it  is  possible  to  derive  a  number  of  relations  between 
loss  and  phase.  One  of  the  simplest  is 


/        /3  du  =  -  (A^  -  AQ)  (53) 

»/ —  »  2 

where  w  =  log  CO/COQ,  /o  ( =  coo/2-n-)  being  an  arbitrary  reference  frequency,  0  is  the  phase 
shift  in  radians,  and  AQ  and  Aw  are  the  losses  in  nepers  at  zero  and  infinite  frequency 
respectively.  This  states  that  the  area  under  the  phase  curve,  when  plotted  on  a 
logarithmic  scale,  depends  only  on  the  difference  in  the  losses  at  zero  and  infinite  frequency. 
A  second  and  possibly  more  useful  relation  is  given  by 


"°°  dA  1  u  1 

—  log  coth  —1  du 
—  co  au  2 


(54) 


where  &Q  represents  the  phase  shift  in  radians  at  any  arbitrary  frequency  /o  (  =  coo/2?r)  and 

u  =  log  W/WQ.    This  result  states  that  the  phase  shift  at  any  frequency  is  proportional  to 

the  derivative  of  the  loss,  on  a 

logarithmic   frequency  scale,   at 

all  frequencies.     It  involves  an 

integration  over  the  entire  fre- 

quency spectrum.     The  function 

log  coth  |  u  |/2  is  in  the  nature 

of  a  weighting  function  and  is 

shown  in  Fig.  17.    Its  value  is 

much  larger  near  the  point  co  —  coo 

and  tends  to  emphasize  the  effect 

of  the  loss  characteristic  in  the 

immediate  vicinity. 

As  an  illustration  of  the  utility 
of  eq.  (54)  let  it  be  supposed  that 
A  —  ku,  which  describes  a  loss 
curve  of  constant  slope  on  a  log- 
arithmic scale  of  20k  db  per 
decade  (Qk  db  per  octave).  The 
associated  phase  shift  is  readily 
found  to  be  k-rr/2  radians.  As  a 


J.O 


10. 


FIG.  17.    Weighting  Function  in  Loss-phase  Formula 


second  example  consider  the  discontinuous  loss  characteristic  of  Fig.  IS.  Here  the  loss  is 
assumed  to  be  zero  below  a  specified  frequency  /o  (  =  coo/2?r)  ,  and  has  a  constant  slope  of 
6k  db  per  octave  above  coo/27r.  The  associated  phase  shift  shown  in  Fig.  18  is  symmetrical 
about  the  frequency  /o,  at  which  point  fi  =  k-jr/4,  and  approaches  the  value  kir/2  radians 
as  frequency  increases.  At  low  frequencies  the  phase  shift  is  substantially  linear  and  is 
giyen  by  jS  =  2&co/(7rcoo). 

Since  the  phase  characteristic  corresponding  to  the  sum  of  two  loss  characteristics  is 
the  sum  of  the  two  phase  characteristics  corresponding  to  the  separate  loss  characteristics, 
it  is  possible  to  add  a  number  of  such  simple  characteristics  together  to  simulate  more 
complicated  loss  characteristics  and  to  evaluate  the  corresponding  phase  shift.  An  ex- 
ample is  furnished  by  Fig.  19,  which  shows  a  phase  curve  derived  as  the  algebraic  sum  of 
three  simple  solutions  of  the  type  shown  in  Fig.  18.  By  proceeding  in  similar  fashion  it  is 
possible  to  derive  the  phase  shift  corresponding  to  almost  any  loss  characteristic  without 
actually  performing  the  integration  indicated  in  eq.  (54).  In  this  connection  it  should  be 
observed  that,  if  both  the  loss  and  the  corresponding  phase  can  be  specified  at  all  fre- 


5-18 


ELECTRIC   CIRCUITS,   LINES,   AND   FIELDS 


quencies,  the  problem  of  designing  an  equalizer  having  the  inverse  characteristics  is  imme- 
diately reduced  to  that  of  finding  a  two-terminal  impedance  for  which  both  resistance  and 
reactance  are  known.  See  article  7. 


llOfci 

Mfofcj 

90k 


70k 
60k 

sofc 

40k 
30k 


20k 


10k 


-lOfc 


'&. 


1.0 


22k 
20k 


I0k% 
8k 


2k 


iOD. 


FIG.  18.    Semi-infinite  Slope  of -.Attenuation  (4)  and  Corresponding  Phase  Shift  (J?) 


25k 
20k 


0 
30k 

m   10fc 

! 

I    ° 

— lOfc 
— 20k 


100. 


FIG.  19.     Phase  Curve  Corresponding  to  Sum  of  Three  Semi-infinite  Attenuation  Slopes 

LOSS  EQUALIZERS.  The  networks  commonly  used  as  loss,  or  attenuation,  equalizers 
=are  shown  in  Fig.  20,  which  also  gives  the  expressions  for  the  insertion  loss  factor. 

The  networks  shown  are  divided  into  classes  based  on  their  transmission  and  impedance 
properties.  The  simple  series  and  shunt  networks  designated  la  and  Ib  are  most  useful 


CORRECTIVE  NETWORKS 

for  simple  problems.    Their  transmission  characteristics  depend,  of  course,  on 
nations. 


5-19 


The  L-type  networks  desig- 
nated as  Ha  and  lib  have  the 
same  form  for  the  insertion 
loss  factor  as  la  and  IZ>.  They 
have  the  additional  property, 
however,  that  the  input  imped- 
ance is  equal  to  a  constant,  -&0, 
when  they  are  terminated  in  RQ 
on  the  output.  Consequently, 
several  sections  can  be  operated 
in  tandem  without  interaction, 
and  the  insertion  loss  of  each 
section  is  independent  of  the 
generator  impedance. 

The  symmetrical  T,  x,  bridged 
T,  and  lattice  networks  shown 
as  Ilia,  Ill&JIIc.and  Hid  have 
insertion  loss  factors  identical  to 
the  preceding  network.  How- 
ever, they  have  constant-resist- 
ance image  impedances.  As  a 
consequence  the  insertion  loss 
factor  has  the  form  shown  if 
either  the  generator  or  load  im- 
pedance has  the  value  RQ.  Net- 
work IIIc  is  the  most  generally 
used  because  it  requires  fewer 
elements  than  any  of  the  others. 
The  network  shown  as  IV  is 
a  general  constant-resistance 
lattice  structure  of  which  IIIo* 
is  a  special  case.  This  is  the 
most  general  form  of  constant- 
resistance  network,  since  any 
transmission  characteristic 
which  can  be  realized  can  be 
obtained  with  a  structure  of 
this  form.  See  article  7.  It  is 
possible,  therefore,  to  base  the 
design  of  all  equalizers  on  this 
structure,  even  though  the  net- 
work may  be  built  in  one  of  the 
other  forms  when  such  a  conver- 
sion leads  to  a  physical  network. 
The  insertion  loss  factor,  ee,  is 
somewhat  more  complicated  for 
this  network  than  for  the  others 
listed. 

In  terms  of  the  admittance  of 
the  Z\  arm 


(55a) 


Configuration 


+  1 


where  YI  =  (^  +  jBi. 

If  61  is  assumed  to  be  con- 
stant with  frequency,  the  net- 
work can  be  shown  to  behave 
as  either  a  minimum  phase  or  a 
non-nodnimum  phase  network  as 


both  termi- 
tnsertlon  Loss  Factor 

•7, 

Ro~Ri+R2 


Ro* 


RiR2 


116 


UTa 


IIT5 


--£•          ZiZ^-Ro 


=!+7 


xv 


FIG.  20.    Equalizer  Configurations  and  Insertion  Loss  Factors 


5-20  ELECTRIC   CIRCUITS;   LINES,   AND  FIELDS 


the  product  RoGi  is  greater  or  less  than  unity.    An  explicit  expression  for  YiRo  in  terms 
of  6  is  evidently 

RoYl  "  "  C0th 


A  fairly  general  approach  to  equalizer  design  based  on  the  constant-resistance  lattice 
structure  has  been  discussed  by  0.  J.  Zobel.*  This  method  provides  a  systematic  means 
of  determining  a  network  to  satisfy  a  given  loss  requirement. 

Referring  to  eq.  (55a)  it  is  a  relatively  simple  matter  to  evaluate  <?  corresponding  to  a 
given  admittance  7i.  A  more  difficult  problem  is  to  determine  7i  for  a  given  set  of 
values  for  a,  since  in  a  typical  problem  the  corresponding  value  of  (3  is  usually  unknown. 

The  design  procedure  is  briefly  as  follows.  The  insertion  power  ratio  is  written  as  a 
function  of  frequency  thus 

=  ao  +  am2  +  02CQ4  +  •  •  • 

bo  +  6iw*  +  52cu4  +  -  -  •  Wj 

where  the  a's  and  b's  are  real  constants.  A  set  of  linear  equations  in  the  a's  and  6's  is 
determined  by  assigning  values  to  620!  at  specific  frequencies.  As  many  frequencies  are 
selected  as  the  number  of  coefficients  to  be  determined.  The  solution  of  this  set  of  equa- 
tions gives  the  values  of  the  a's  and  6's  required,  and  the  function  e2*  is  determined.  The 
roots  of  the  numerator  and  denominator  of  e2a  are  then  found  in  terms  of  p2  =  —  co2. 

Since  the  function  must  lead  to  a  physical  network  in  order  that  it  be  useful  the  condi- 
tions must  be  satisfied  that 

1.  eza  be  greater  than  1.0  for  all  values  of  co. 

2.  The  roots  of  the  numerator  may  not  occur  at  real  frequencies  —  otherwise  e2a  would 
have  a  point  of  infinite  gain. 

3.  The  roots  may  be  real  or  conjugate  complex  pairs. 

4.  The  roots  of  the  denominator  may  be  anywhere  in  the  p2  plane,  but  those  that  fall 
on  the  negative  real  axis  must  be  of  even  multiplicity  —  otherwise  e2a  will  jump  from 
4-  oo  to  —  oo  in  this  region,  and  so  violate  restriction  1. 

If  these  conditions  are  satisfied  the  design  proceeds.  If  not,  a  change  is  required; 
either  the  assumed  form  for  tza  or  the  matching  points  must  be  altered. 

To  specify  the  network  the  function  ee  is  formed.  The  roots  of  e2*  are  known  in  terms 
of  p2,  and  the  corresponding  roots  in  terms  of  p  are  readily  available,  since  they  occur 
in  +  and  —  pairs. 

The  roots  of  the  numerator  used  to  form  e0  must  be  in  the  left-half  of  the  p  plane.  The 
roots  of  the  denominator  are  not  necessarily  so  restricted.  However,  if  this  restriction  is 
applied  to  them  also,  the  solution  for  ed  will  lead  to  a  minimum  phase  network. 

Utilizing  the  resulting  function  e6  the  lattice  arm  impedance  Zi,  or  admittance  FI,  is 
determined  from  eq.  (556).  The  network  which  exhibits  this  impedance  may  be  found 
in  a  variety  of  ways.  A  general  method  for  solving  this  problem  has  been  described  by 
O.  Brune.f 

As  an  example  of  the  design  process  consider  that  the  function  to  be  matched  is 

#*  =  1  +  w6 
In  terms  of  p2  —  —  o>2  this  becomes 

<?«  =  1  -  p6 

The  roots  of  e20£  are  given  by  p2  =  1.0  and  p2  =  —  0.5  =t  ^0.866.  In  terms  of  p  the  roots 
having  negative  real  parts  are  then  p  =  —  1  and  p  =  —  0.5  db  j'0.866,  and 

€0  =  (p  -f  0.5  -  y0.866)(p  +  0.5  +  y0.866)(p  +  1) 

=  p3  +  2P2  +  2p  4-  1 
whence 

' 


Expanding  this  as  a  continued  fraction  by  removing  alternately  poles  and  zeros  the  result 
is  obtained  that 


The  corresponding  impedance  of  the  series  arm  of  the  lattice  is  shown  in  Fig.  21  for  unit 
impedance  and  unit  frequency. 


*O.  J.  Zobel,  Distortion  Correction  in  Electrical  Circuits  with  Constant  Resistance  Networks, 

.S.T.J.,  Vol.  VII,  July  1928,  pp.  438-534. 

t  Journal  of  Mathematics  and  Physics,  M.I.T.,  VoL  X,  October  1931. 


COKEECTIVE   NETWORKS  5-21 

For  many  problems  the  analytic  method  of  design  which  has  been  discussed  is  cumber- 
some and  unsuitable.  For  example,  if  it  be  required  to  equalize  a  measured  characteristic 
with  only  a  fair  degree  of  accuracy,  the  effort  required  to  obtain  a  precise  solution  is  not 
justified.  In  such  cases  a  knowledge  of  the  behavior  of  the  simpler  forms  of  two-terminal 
impedances  can  be  usefully  applied.  The  ability  to  visualize  the  frequency  characteristic 
of  a  configuration  of  coils,  condensers,  and  resistances  is  an  essential  part  of  the  designer's 
equipment. 

Probably  the  most  useful  configuration  for  design  purposes,  since  it  is  one  of  the  simplest, 
is  that  shown  as  IIIc  in  Fig.  20,  in  which  the  impedance  Z\ 
consists  of  a  reactance  shunted  by  a  resistance.    For  this  % 

network  o— 


-  1  +  .    v  (57) 


where  GI  is  a  constant.    Maximum  loss  occurs  when  BI  —  0,     .-,      01      T   ,.  .      ,        T 
while  minimum  loss   (a  =  0)  occurs  when  BI  -»  «>.     The     FlG'  21>     Lg££  ^ 

design  problem  is  reduced,  once  a  selection  has  been  made  of 

the  maximum  loss  required,  to  finding  a  suitable  reactance,  or  susceptance,  to  match  the 
loss  curve  over  the  required  interval.    From  eq.  (57)  it  is  evident  that 


(59) 


Bf) 

whence 


For  known  values  of  e2"  corresponding  values  of  BI  can  be  found.  A  relatively  slight 
effort  is  required  to  determine  the  required  susceptance.  The  conditions  for  physical 
readability  are  here  merely  that  the  zeros  and  poles  occur  at  real  frequencies,  that  they 
alternate,  and  that  the  reactance  function  behave  as  a  simple  coil  or  condenser  at  zero  or 
infinite  frequency. 

In  applying  this  process  to  build  up  complicated  loss  frequency  characteristics  as  the 
sum  of  several  equalizer  sections  the  skill  of  the  designer  is  evidenced  by  his  ability  to 
select  easily  realizable  characteristics  for  the  component  sections. 

PHASE  EQUALIZERS.  The  constant-resistance  lattice  network  IV  of  Fig.  20  be- 
becomes  an  "all-pass"  network  when  the  impedances  Zi  and  Zz  are  specified  as  pure 
reactances.  Then 


Ro-  JX1 

e2«  =1.0     and     tan  f  =  =fc  ^ 
2  ti 

There  are  two  basic  networks  of  this  type,  which  are  shown  in  Fig.  22.    For  the  first  of 
these  (Type  I) 

9  _  RQ  +  Lp  _       p  +  BO/L 


~  R0-Lp  -        p  - 

The  zero  of  this  expression  is  real  and  negative,  and  the  pole  is  real  and  positive.    In  the 
p  plane  they  occur  symmetrically  on  either  side  of  the  origin.    The  phase  curve  correspond- 
ing to  this  network  is  shown  in  Fig.  22;  the  critical  frequency  co0 
In  the  more  complicated  network  shown  as  Type  II  in  Fig.  22 


where 


and 


<*\  i  r 

PI  -  -»  =  T MC^  [- 


5-22 


ELECTRIC  CIRCUITS,  LINES,  AND  FIELDS 


When  4:R(?C/L  :g  1.0  the  zeros  and  poles  are  real.  The  zeros  occur  in  the  left  half  of  the 
p  plane,  and  the  poles  are  in  the  right  half-plane.  The  network  is,  therefore,  equivalent 
to  two  of  the  simple  types  in  tandem. 

When  4:R<?C/L  >  1.0  the  zeros  and  poles  are  complex.  Again  the  zeros  are  in  the  left 
half-plane  and  are  conjugate  complex  numbers.  The  poles  are  again  in  the  right  half- 
plane  and  are  symmetrically  disposed  about  the  origin.  As  would  be  expected  a  wide 
variety  of  phase  characteristics  can  be  obtained.  Some  typical  curves  are  shown  in  Fig.  22 
plotted  against  w/co0.  Note  that,  when  4RQZC/L  =  1.0,  the  phase  shift  is  exactly  twice 
that  of  Type  I. 

Increasing  the  complexity  of  the  reactances  in  the  lattice  arms  merely  introduces  addi- 
tional zeros  and  poles  in  the  expression  for  ee.  If  the  network  is  to  continue  to  be  an  all- 
pass  network,  the  zeros  and  poles  must  occur  in  pairs  having  characteristics  similar  to 


FIG.  22.    Typical  Phase  Characteristics  of  All-pass  Networks 

those  noted  above.  Consequently,  it  is  possible  to  break  down  a  more  extensive  set  of 
zeros  and  poles  into  groups,  each  group  corresponding  to  a  physical  network  of  either 
Type  I  or  Type  II.  A  number  of  these  simpler  networks  in  tandem  will  provide  charac- 
teristics exactly  similar  to  those  of  the  more  complicated  lattice. 

BIBLIOGRAPHY 

Bode,  H.  WM  Network  Analysis  and  Feedback  Amplifier  Design.    D.  Van  Nostrand  (1945). 
Gardner  and  Barnes,  Transients  in  Linear  Systems,  Vol.  I. 

Bmne,  O.,  Journal  of  Mathematics  and  Physics,  M.I.T.,  Vol.  X,  October  1931,  pp.  191-235. 
Darlington,  S.,  Journal  of  Mathematics  and  Physics,  M.I.T.,  Vol.  XVIII,  No.  4,  September  1939,  pp. 

257-353. 

Foster,  R.  M.f  A  Reactance  Theorem,  B.S.T.J.,  April  1924,  pp.  259-267. 
Foster,  R.  M.,  Geometrical  Circuits  of  Electrical  Networks,  Trans.  A.I.E.E.,  June  1932. 
Guillemin,  E.  A.,  Communication  Networks,  Vol.  II. 
Zobel,  O.  J.,  Distortion  Correction  in  Electrical  Circuits,  B.S.T.J.,  Vol.  VII,  July  1928,  pp.  438-534. 


RECURRENT  NETWORKS 

By  P.  H.  Richardson 

Early  work  on  transmission  networks  was  from  the  viewpoint  of  wave  propagation  in 
uniform  media.  Later  work  introduced  the  methods  of  particle  dynamics,  and  networks 
were  treated  as  vibrating  systems.  In  treating  the  problem  of  recurrent  networks,  that 
is,  the  problem  of  a  number  of  similar  networks  in  tandem,  the  terminology  of  the  earlier 
method  is  most  useful.  A3  might  be  expected,  the  problem  is  readily  handled  in  terms  of 
equivalent  line  parameters  such  as  the  image  parameters  previously  defined.* 

*  Solutions  of  the  general  problem  in  terms  of  botk  image  and  iterative  parameters  are  given  by 
E.  A.  Guillemin,  Communication  Networks,  Vol.  II,  pp.  163-175.  In  the  case  of  symmetrical  networks 
the  two  sets  of  parameters  are  identical. 


SYMMETRICAL  NETWORKS 


5-23 


11.  SYMMETRICAL  NETWORKS 

CURRENT  AND  VOLTAGE  RELATIONS.  Consider  the  tandem  combination  of  two 
symmetrical  networks  shown  in  Fig.  1.  The  image  impedance  is  assumed  to  be  the  same 
for  both  structures,  while  the  transfer  constants  QI  and  02  are  different.  To  find  the  voltage 


FIG.  1.    Tandem  Combination  of  Symmetrical  Networks 

at  the  junction  x-x'  replace  the  generator  and  network  to  the  left  of  the  junction  by 
an  equivalent  Thevenin.  generator.    Then  the  open  circuit  voltage  at  x-x' 


EZi 


The  voltages  Ex  and  EI  are  evidently  given  by 

#.._ 

,_  „.      and 


E 


The  impedances  Zi',  Zs',  and  J£x*  are  given  by  the  relations 

Zi'-. 


(i) 


(2) 


(So) 


(36) 


(3c) 


where  pi  ~     *    ,    -    and  p2  =     *   .    ,,    are  the  reflection  coefficients  at  junctions    1-1' 

Zti  +  Zil  42  -\-  Z,I 

and  2-2'  respectively. 

Substituting  in  eq.  (2)  It  can  be  demonstrated  that 


where 


jfia;          £ri 

J'  "  2?  "  Zi 


and 


(4) 
(5) 

(6) 


Note  that  AI  and  5i  depend  only  on  the  sum  of  the  two  transfer  constants  and  the  reflec- 
tion coefficient  at  the  output  junction. 

INCIDENT  AND  REFLECTED  WAVES.  The  term  JM^-ft  is  called  the  incident 
component  of  the  voltage,  and  EiBi€6i  is  called  the  reflected  component  of  the  voltage.  Like- 
wise the  terms  (Ei/Zi)Aie~8i  and  —(Ei/Zi)B^  are  called  the  incident  and  reflected 
components  respectively  of  the  current  at  the  junction  x-xf.  Note  that  if  Zz  —  Zi, 
that  is,  if  the  impedances  are  matched  at  the  output,  there  is  no  reflected  component  since 
BI  =  0.  Also,  if  61  +  0$  has  a  large  positive  real  part,  BI  becomes  very  small,  and  again 
the  reflected  component  vanishes.  The  voltage  ratio  and  the  current  ratio  are  both  equal 
to  €~dl,  when  the  reflected  "wave"  is  zero. 

IMPEDANCE  RELATIONS.  The  pair  of  networks  having  like  image  impedances 
behave  as  regards  the  input  and  output  meshes  as  though  they  together  constituted  one 
network  of  image  impedance  Zi  and  transfer  constant  0i  +  #2-  When  the  output  im- 
pedance matches  Zi,  pa  =  0  and  the  impedances  Z\  and  Zx"  are  both  equal  to  Zi.  Note 
also,  if  the  real  part  of  0i  +  62  is  large,  the  input-impedance  again  is  equal  to  Zi. 


5-24 


ELECTRIC  CIRCUITS,  LINES,   AND  FIELDS 


A  relation  which  is  frequently  useful  in  design  problems  can  be  obtained  from  eq.  (3). 
The  reflection  coefficient  at  the  input  terminals 


p./ 

pl 


l'  -  Zl 


Thus  the  reflection  coefficient  at  the  input  terminals,  pi/,  is  simply  related  to  the  reflection 
coefficient  at  the  output  terminals.  If  0i  +  Q%  is  a  pure  imaginary,  then  |  p\'  \  =  |  p2  | 
Also,  if  a  is  the  real  part  of  0i  -f-  62,  then  |  pi'  |  —  1  P2  I 


*-2os 


12.  UNIFORM  LINES—  NETWORKS  WITH  DISTRIBUTED  CONSTANTS 

When  the  physical  dimensions  of  a  network  are  comparable  to  the  wavelength  of  the 
electric  current  flowing  in  it,  account  must  be  taken  of  the  fact  that  the  series  resistance 
and  inductance  of  each  wire,  and  the  shunt  capacitance  and  leakage  between  wires,  are 
"distributed."  Such  networks  are  usually  termed  transmission  lines  (see  also  Section  14 
in  volume  on  Electric  Power),  but  in  the  case  of  very  high-frequency  currents  (5  meters 
or  less)  a  network  contained  within  an  ordinary  room  may  have  to  be  similarly  treated. 

The  usual  transmission  line  in  communication  circuits  consists  of  two  similar  parallel 
wires,  or  a  single  wire  enclosed  in  a  conducting  cylindrical  sheath.  At  every  point  on  the 
line,  some  current  flows  from  one  wire  to  the  other,  or  to  the  sheath,  owing  to  capacitance 
and  leakage  conductance  resulting  from  the  imperfect  dielectric  between  them.  In  conse- 
quence of  this,  the  current  in  each  conductor  varies  along  the  line.  This  is  illustrated  in 
Pig.  2,  which  shows  an  elementary  section  of  a  balanced  line,  or  of  a  completely  unbalanced 


%Zdx 


%Zdx 


o  —  VV^-  nfflT^--nSW^-AM  —  oC  Ao  —  -W  —  ''TO^-o—  'TKftP  —  W>  —  o 


>Y  dx^p 


%z  dx 


%z  dx 


-dx- 


E 
-dx- 


where  Z-r+jLco  and  Y=£7+j"Cw 
FIG.  2.    Elementary  Section  of  a  Transmission  Line 

line  equivalent  to  the  coaxial  type  of  construction.  The  current  entering  at  A  is  not  the 
same  as  that  leaving  at  (7,  owing  to  the  capacitance  and  conductance  shunted  between 
B  and  E.  Hence  the  total  drop  due  to  the  total  resistance  of  the  line  is  not  this  resistance 
multiplied  by  the  current  at  any  point  on  the  line.  Simple  impedance  equations  cannot 
be  written;  simple  differential  equations  can  be  given  (for  this  solution  see  Section  14  in 
volume  on  Electric  Power) ,  and  these  offer  one  method  of  attack. 

An  alternative  method  is  based  on  the  assumption  that  such  a  uniform  transmission 
line  may  be  considered  to  be  composed  of  an  infinite  number  of  symmetrical  networks, 
each  of  which  corresponds  to  an  infinitesimal  length  of  line.  If  r  is  the  resistance  and  L 
the  inductance  per  unit  length  (for  two  wires,  sometimes  called  a  loop-mile  when  the 
unit  of  length  is  a  mile) ,  and  g  the  conductance  and  C  the  capacitance  between  the  wires, 
or  from  the  central  conductor  to  the  sheath,  per  unit  of  length,  then,  in  a  section  of  length 
dx,  there  will  be  a  resistance  r  dx,  an  inductance  Z/  dx,  conductance  g  dx,  and  capacitance 
C  dx.  As  dx  approaches  zero  the  sections  become  smaller,  and  a  line  of  these  sections 
approaches  a  line  with  uniformly  distributed  constants. 

VOLTAGE  AND  CURRENT  RELATIONS.    Let  Z  =  T  +  jLu  =  |  Z  \  /J^  be  the  series 
impedance  per  unit  length,  and  let  Y  =  g  +  yCco  =  |  Y  \  /6y 

A  A  be  the  shunt  admittance  per  unit  length  of  line.    Then 

for  the  T  network  equivalent  to  an  infinitesimal  length 
of  line  (see  Fig.  3)  for  which  6  =  d& 


PIG.  3.     T  Network  Equivalent  to 
Smooth  Line 


Zc  =  T~  ~ 

dd 


- 
Y  dx 


(8a) 


(86) 


UNIFORM  LINES  5-25 

Thus 

7  -  \ 

Zl  - 


This  impedance  is  frequently  called  the  characteristic  impedance  of  the  uniform  line;  the 
terms  image  impedance  and  iterative  impedance  which  are  also  used  are  somewhat  more 
clearly  denned.    They  are,  of  course,  identical  to  one  another  for  the  symmetrical  lint*. 
From  eqs.  (8)  it  follows  that  d6  =  VZT  dx,  whence 


(10) 

where  x  represents  the  distance  to  a  point  on  the  line  from  the  sending  end.  The  quantity 
7  is  called  the  propagation  constant  of  the  line  and  is  evaluated  per  unit  length  of  line.  For 
a  given  length  of  line  the  total  propagation  constant  is  seen  to  be  identical  to  the  image 
transfer  constant  of  the  equivalent  symmetrical  network.  7  is  a  complex  number  and  can 
be  expressed  as  y  =  a  +  jjS,  where  «  and  j3  are  real.  Expanding  eq.  (10)  and  separating 
reals  and  imaginaries 


+  (rg  -  LCco3)]  -(Ha) 


L2a>2)(g2  +  C2^)  -  (rg  - 

The  parameters  a.  and  $  are  called  the  attenuation  constant  and  phase  shift  constant,  re- 
spectively, of  the  line.  Note  that  they  are  both  functions  of  frequency. 

The  relations  given  by  eqs.  (9)  and  (10)  indicate  the  similarity  between  the  behavior  of 
networks  having  lumped  constants  and  the  behavior  of  uniform  lines.  At  some  point  a 
distance  x  from  the  sending  end  of  a  line  of  length  I  the  voltage  Ex  and  current  Ix  are 
given  by  eqs.  (4),  (5),  and  (6)  of  article  1,  where  0i  =  yx  and  82  —  *y(l  —  x).  Similarly 
the  input  impedance  of  the  line  is  given  by  eq.  (3a)  . 

INCIDENT  AND  REFLECTED  WAVES.  The  instantaneous  voltage  at  any  point 
on  the  wire  is 


and 

ix  =  Real  part 


ex  =  Real  part  [E^^A^-v*  + 

=  |  Si  |  [I  Ai  |  €-«*  cos  M  -  |8a:  +  84)  +  |  Bi  |  e**  cos  (tat  +  0x  +  5*)3        (12) 

|~  J^  4**  (4ie—  >*  - 

1  €~^  cos  (w*  -  jSa;  +  6^1  -  *)  -  I  Bi  |  «aa;  cos  M  4-  ]5a;  +  SB  -  *)]      (13) 
where  J.i  =  |  Ai  \  /5A,  BI  =  |  BI  ]  /5g,  Zj  =  |  Z/  |  /^  and  EI  is  real. 

Each  of  these  expressions  is  composed  of  two  components,  the  incident  "wav&"  which 
decreases  in  magnitude  as  x  increases,  and  the  reflected  "wave"  which  increases  as  x  increases. 
These  components  are  called  "waves"  because  they  appear  to  travel  along  the  wire  with 
a  velocity  v  =  co/£.  The  distance  between  two  consecutive  points  on  the  line  at  which 
cos  (u>t  —  fix  +  5  A)  and  its  derivative  have  the  same  algebraic  values  for  a  fixed  value  of  t 
is  called  a  wavelength  (X),  so  that  /3A  =  2-Tr.  The  time  elapsing  during  a  complete  cycle 
of  values  is  called  the  period  (T)  ,  so  that  T  —  I//.  In  terms  of  these  values  the  velocity 
of  phase  propagation  of  the  waves  is  T?  =  <u/£  ==  \/T. 

STANDING  WAVES.  The  combination  of  the  incident  and  reflected  waves,  adding 
sometimes  in  and  sometimes  out  of  phase,  causes  variation  in  the  value  of  the  voltage 
and  current  with  position  along  the  line.  The  expression  for  the  amplitude  of  the  voltage  is 


I  Ex  |   =  |  Ei  |  V|  Ai  |26-2o*  +  |  jgj  |2e2oz  +  2|  AI  ]|  BI  |  COS  (203  +  65  -  8 A)      (14) 

and  for  the  current, 

|  Ix  I  =    -~    V|  -di  |2€-2o^  -j-  |  j5x  |2€2o:a:_  2J  AI  ||  BI  |  cos  (2j3x  +  SB  —  5j.)       (15) 

each  of  which  will  have  maximum  and  minimum  values  along  the  line  whenever  |  BI  \  ?£  0, 
since  cos  (2px  +  SB  —  8 A)  changes  more  rapidly  than  e2ax.  When  attenuation  is  negligible 
and  I  Ai  \  =  \  Bi  | 

\EX\  =  |  EI  |[  Ai  1  V2[l  +  cos  (2$x  +  SB  —  8  A)]  (16) 


5-26  ELECTRIC  CIRCUITS,  LINES,   AND  FIELDS 

From  this  it  is  evident  that  there  are  positions  where  \  Ex\  —  2\  E\  \\  A\  \  called  voltage 
loops,  and  others  where  |  Ex  \  —  0  called  voltage  nodes.  Similarly  there  are  current  loops 
and  nodes,  the  current  nodes  corresponding  to  voltage  loops,  and  vice  versa. 

It  should  be  noted  that  the  condition  that  |  AI  \  =  |  BI  \  requires  that  p^~^1  =  ±1.0. 
Thus  true  standing  waves  occur  only  when  the  line  is  a  multiple  of  a  quarter  wavelength 
and  when  pz  —  =1=1.0,  that  is,  when  Zz  is  either  zero  or  infinite. 

INPUT  IMPEDANCE.  The  input  impedance  of  a  uniform  line  terminated  in  an  im- 
pedance #2  at  a  distance  x  from  the  sending  end  is  given  by 

"      z.    1 

l~    ' 


where  pz  —  ^  -  —  •    li  Z%  =  Zi,  then  p2  =  0  and  Zi  =  Zi*   The  variation  of  Zi  with 
£2  -h  Zi 

frequency  is  a  smooth  wave  which  approaches  the  value  VL/C  as  /  increases. 

When  Z%  j£  Zi  the  Zt-  vs.  /  curve  has  maxima  and  minima  which  can  be  approximately 
located.  If  it  be  assumed  that  Zi  and  Z%  are  pure  resistances,  which  together  with  a  are 
independent  of  frequency,  then  the  locus  of  Zi  in  the  Zi  plane  is  a  circle  of  radius 


(18a) 
with  its  center  at 

/I   A.  n^~4ax\         1  ^ 

If  p2  is  positive  (Z%  >  Zi),  Zi  will  have  its  maximum  or  minimum  value  when  e~^x  is  +1 
or  —  1.  This  requires  that  2/3x  =  2mr,  or  (2n  —  I)TT,  where  n  is  any  integer.  Since 
j3  =  CO/B  the  frequency  at  which  the  maximum  or  minimum  occurs  is  given  by 

run  (C)""  —  1^" 

/max.  —  r~     or     /min.  — 


If  /  =  /i  when  n  —  n\  and  /  =  /2  when  n  =  n\  +  1,  then 


2(/2  -  /i) 

But  since  /2  —  /i  is  the  frequency  interval  between  two  successive  maxima,  or  two  suc- 
cessive minima,  the  distance  x  to  the  point  on  the  line  at  which  there  is  an  impedance 
discontinuity  can  be  determined  approximately.  It  should  be  noted  that  the  expressions 
for  the  frequencies  at  which  maxima  and  minima  occur  will  be  interchanged  if  p2  is  negative 
(Z*  <Zfi. 

DISTORTIONLESS  LINES.  In  communication  systems  many  frequency  components 
are  usually  present,  and  it  is  desirable  to  have  uniform  attenuation  with  frequency.  This 
condition  exists  on  a  line  if  r/L  =  g/C,  in  which  case 


a  =  Vrg    and 


both  of  which  are  independent  of  frequency.    Similarly,  j3  —  ccVZC,  which  is  the  condition 
for  linear  phase  shift  and  no  delay  distortion.    In  this  case  also 


ft     VLC 

which  states  that  all  waves  are  propagated  with  the  same  velocity. 

TRANSIENTS  IN  NETWORKS 

By  Harold  A.  Wheeler 

13.  TRANSIENT  DISTURBANCES 

PROPERTIES  OF  TRANSIENTS.  A  transient  disturbance,  in  its  simplest  concept, 
is  one  that  occurs  in  a  time  interval  separated  from  other  disturbances.  In  general,  a 
transient  may  be  superimposed  on  other  transients  or  continuous  waves,  according  to  the 
superposition  theorem,  while  otherwise  retaining  its  own  characteristics.  A  single  tran- 


TRANSIENT  DISTURBANCES 


5-27 


ents,    -§A 

K&).    I 
itch-    I' 


sient  cannot  be  a  periodic  wave  in  the  strict  sense,  although  it  may  be  a  damped  oscillation 
of  a  definite  period.  A  borderline  case  is  the  "periodic  transient,"  which  is  a  periodic  non- 
overlapping  series  of  transients;  each  transient  retains  the  properties  of  a  transient  while 
the  series  has  the  properties  of  a  periodic  wave. 

Any  periodic  wave  can  be  analyzed  into  a  "Fourier  series"  which  is  a  sum  of  sinusoidal 
components  of  frequencies  hi  harmonic  relation.  The  corresponding  representation  of  a 
transient  disturbance  is  possible  by  the  "Fourier  integral,"  which  does  not  give  a  number 
of  distinct  components  but  rather  the  distribution  of  energy  over  the  frequency  spectrum. 
The  Fourier  series  and  the  Fourier  integral  are  analogous  to  the  line  spectrum  and  the 
band  spectrum  in  light  waves. 

In  general,  an  exact  representation  by  a  Fourier  series  requires  an  infinite  number  of 
components  extending  over  the  entire  frequency  spectrum,  but  a  practical  approximation 
requires  only  a  limited  number  of  components  within  a  limited  bandwidth.  The  same 
is  true  with  respect  to  the  limited  bandwidth  required  for  the  practical  reproduction  of  a 
transient.  It  is  essential  to  consider  the  degree  of  approximation  required  or  attained  in 
any  particular  case,  and  to  realize  that  both  the  duration  of  the  transient  and  the  frequency 
bandwidth  are  theoretically  unlimited  although  practically  limited  by  the  sensitivity  of 
the  system. 

Refer  to  Section  9,  Pulse  Techniques,  for  much  information  on  transients  as  exemplified 
by  pulses,  with  emphasis  on  their  application  in  practical  systems. 

TYPES  OF  TRANSIENTS.  It  is  convenient  to  define  transient  disturbances  of  several 
idealized  types.  An  actual  transient  may  be  classified  by  its  similarity  to  one  of  these 
types,  or  as  the  response  of 

a  certain  network  to  one  of  I 

these  types.  ! 

The  unit  step  is  one  of 
the  elementary  transients, 
shown  in  Fig.  l(a)  and  ,  . 

It  may  be  caused  by  switch-    | '  +  —• 

ing  or  keying  a  current  or 
voltage  of  unit  amplitude. 
The  form  of  Fig.  1  (a)  is  also 
called  the  "Heaviside  unit 
function, ' '  with  a  j  ump  from 
zero  to  one.  The  form  (6) 

has  a  jump  from  — 1/2  to  -M/2;  it  is  preferred  for  some  analytical  purposes  because  it  has 
zero  d-c  component.  Campbell  and  Foster  (references  7  and  8)  designate  the  unit  step  as 
jS_i,  called  the  singularity  function  of  —1  order. 

The  unit  impulse,  shown  in  Fig.  l(c),  is  unique  among  transients  in  that  it  has  a  uniform 
frequency  spectrum.  It  is  defined  as  the  derivative  or  slope  of  the  unit  step;  therefore  it 
has  unit  area  as  the  product  of  very  large  amplitude  and  very  small  duration.  It  is  also 
called  by  Hansen  the  "delta  function"  (reference  38).  Campbell  and  Foster  (references 
7  and  8)  designate  the  unit  impulse  as  So,  called  the  singularity  function  of  zero  order. 

In  practice,  the  step  is  easier  to  generate  because  of  its  limited  amplitude,  and  the 
transient  response  thereto  has  limited  amplitude.  The  impulse  may  be  approximated 
with  limited  amplitude  if  its  duration  is  reduced  until  its  frequency  spectrum  is  substan- 
tially uniform  over  a  frequency  bandwidth  sufficient  for  any  particular  tests.  Most  net- 
works have  some  integrating  action,  so  their  response,  even  to  an  ideal  impulse,  would 
have  limited  amplitude. 

Some  oscillatory  transients  are  illustrated  in  Fig.  2.  In  general,  there  are  reversals  of 
polarity,  which  may  or  may  not  be  periodic  in  time.  The  transient  response  of  a  practical 

network  is  usually  oscilla- 
tory in  some  degree.  Fig- 
ure 2(o)  shows  a  damped 
oscillation  which  is  a  com- 
mon occurrence;  Fig.  2(6) 
shows  a  pulse-modulated 
wave  including  several 

f   ^  _.  cycles  of  a  carrier  wave, 

W  W  (C)  while  (c)  shows  a  carrier 

wave  with  a  step  in  its 
modulation  envelope.  A 


(a) 


(o)s, 


FIG.  1.    Unit  Step  and  Unit  Impulse 


FIG.  2.    Oscillatory  Transients 


carrier  wave  cannot  be  conceived  as  modulated  by  the  ideal  impulse,  because  the  duration 
is  insufficient  to  retain  the  identity  of  the  carrier  frequency. 

Practical  steps  have  finite  slope,  and  practical  pulses  have  finite  duration  and  amplitude. 


5-28 


ELECTRIC   CIRCUITS,   LINES,   AND  FIELDS 


For  test  purposes,  however,  both  can  be  made  to  approach  the  ideal  as  closely  as  required, 
so  the  transient  response  of  a  network  to  the  ideal  types  can  be  tested. 

A  pulse  to  be  used  for  modulating  a  carrier  wave  is  sometimes  denoted  a  "d-c'*  pulse 
to  distinguish  it  from  the  modulated  carrier  of  pulse  envelope.  The  essential  difference 
is  in  the  frequency  spectrum,  the  former  having  most  of  its  energy  concentrated  in  a  band 
including  zero  frequency,  and  the  latter  in  a  band  including  the  relatively  high  carrier 
frequency.  The  direct  and  carrier  pulses  are  distinct  concepts  if  their  respective  frequency 
bands  are  separate. 

Amplitude  modulation  of  a  carrier  wave  generates  symmetrical  sidebands,  and  pulse 
modulation  follows  this  rule.  The  carrier  and  both  sidebands  are  needed  for  exact  repro- 
duction of  the  modulation.  A  small  modulation  superimposed  on  a  continuous  carrier, 
as  illustrated  in  Fig.  2(c),  can  be  approximately  reproduced  in  a  system  responsive  to  one 
sideband  and  one-half  the  relative  carrier  amplitude,  as  employed  in  television  (references 
5,  11,  12,  16-19,  21,  24,  25,  and  Section  20). 

FREQUENCY  SPECTRUM.  Figure  3  shows  the  frequency  spectrum  of  several  kinds 
of  disturbances.  The  pure  sine  wave  has  a  single  frequency  component  (a).  A  periodic 
wave  in  general,  such  as  a  repeating  short  pulse,  has  a  series  of  frequency  components  of 

relative  amplitude  indicated 
by  an  envelope  of  the  lines 
(6) .  A  transient  has  a  con- 
tinuous frequency  spectrum 
(c) .  If  the  transient  has  the 
same  shape  as  a  single  cycle 
of  the  periodic  wave,  the  con- 
tinuous spectrum  (c)  has  the 
same  frequency  distribution 
as  the  envelope  of  lines  (6) . 
a  fundamental  frequency,  denoted  /i  in  Fig.  3,  and  all 


~l 


Frequency 


fa 


FIG.  3.    The  Frequency  Spectrum  of  Periodic  Waves  and  Aperiodic 
Disturbances 


Periodic  disturbances  have  .     , 

components  are  harmonically  related  on  integral  multiples  of  this  frequency.  A  transient 
lacks  a  fundamental  frequency  but,  like  the  periodic  wave,  still  requires  frequency  com- 
ponents over  a  certain  band  width  for  its  reproduction  with  sufficient  approximation.  In 
Fig.  3(c),  a  nominal  cutoff  frequency  fe  may  be  defined  in  such  a  way  as  to  include  most 
of  the  energy  of  the  frequency  spectrum.  For  pulse  operation,  the  nominal  cutoff  fre- 
quency on  the  frequency  spectrum  of  amplitude  may  be  denned  as  the  boundary  of  a 
rectangle  (drawn  as  shown  in  dotted  lines)  having  the  same  area  as  the  amplitude  spectrum 
(reference  11). 

In  the  line  spectrum  of  a  periodic  wave,  each  component  has  a  definite  amplitude  which 
may  be  expressed  in  terms  of  current  or  voltage  (or  analogous  linear  quantities)  .  In  the 
band  spectrum  of  a  transient,  however,  the  amplitude  is  expressed  per  unit  of  frequency 
bandwidth,  in  units  such  as  the  volt  per  cycle  per  second  or  volt-second.  This  concept 
is  elusive,  but  the  shape  of  the  spectrum  indicates  clearly  the  relative  importance  of  the 
various  frequency  components. 

As  an  alternative,  the  frequency  spectrum  may  be  presented  in  terms  of  relative  energy 
instead  of  amplitude.  It  is  then  expressed  in  terms  of  energy  per  unit  of  frequency  band- 
width, in  units  such  as  the  joule  per  cycle  per  second  or  joule-second.  The  area  under  the 
curve  is  the  total  energy  of  the  transient. 

The  frequency  spectrum  of  Fig.  3  shows  the  relative  response  of  a  receiver  of  very  narrow 
and  constant  bandwidth  as  it  is  tuned  over  the  frequency  range.    Spectrum  analyzers  are 
operated  on  this  principle,  some  of  which  show  the 
spectrum  directly  on  an  oscilloscope  (Section  9). 

SPEED  OF  INFORMATION.  In  communica- 
tion of  any  kind,  the  available  frequency  band- 
width is  limited,  and  this  may  restrict  the  speed  of 
transmission  (references  2,  4-6,  9-11,  and  Section 
9)  .  This  is  one  of  the  principal  limitations  in  pic- 
ture  transmission. 

As  a  simple  example,  Fig.  4  shows  the  code  pulses 
for  the  word  "as."  The  pulse  pattern  (a)  contains 
dots  and  dashes  in  the  form  of  short  and  long 
pulses.  It  is  necessary  to  distinguish  the  presence 
or  absence  of  a  pulse  at  intervals  of  one  pulse  width. 


A     A     A 

^  --  \  ---  /\7\T\  -- 
*  -  '    *  —  *'    * 


(&) 

FIG.  4.     Speed  of  Information  by  Pulses 


Increasing  the  speed  or  decreasing  the  frequency  bandwidth  up  to  a  certain  limit  has  the 
effect  of  rounding  the  pulses  (6)  but  not  filling  in  the  space  between  pulses.  This  much 
distortion  is  permissible  and  economical  in  practice. 

The  rounded  pulses  in  Fig.  4(6)  have  a  frequency  spectrum  similar  to  that  in  Fig.  (3c) 


BEHAVIOR  OF  NETWORKS 


5-29 


if  the  dots  have  a  width  2tc  related  to  the  nominal  cutoff  frequency  fe  as  follows  (reference 
11): 

2tc  "  W  (1) 

4/C 

This  means  that  the  shortest  pulse  or  space  has  a  duration  of  1/2  cycle  at  the  nominal 
cutoff  frequency.  The  speed  of  information  is  proportional  to  the  number  of  pulses  in  a 
given  interval  which  is  proportional  to  the  frequency  bandwidth. 

A  completely  modulated  carrier  requires  twice  the  bandwidth  because  double-sideband 
operation  is  essential,  A  partially  modulated  carrier,  with  single-sideband  operation  as  in 
television,  requires  only  slightly  more  bandwidth  than  the  modulating  pulses  but  also 
requires  more  power  to  surmount  background  noise  and  interference. 


14.  BEHAVIOR  OF  NETWORKS 

DIFFERENTIATION  AND  INTEGRATION.  A  linear  network  operates  only  on  the 
amplitude  and  phase  of  a  sine  wave,  retaining  the  wave  form.  Though  the  same  is  true 
of  each  component  of  a  transient,  the  operation  on  all  components  may  greatly  change 
its  shape.  The  simplest  distorting  operations  of  networks  are  differentiation  and  inte- 
gration. 

The  two  basic  differentiating  networks  are  shown  in  Fig.  5.  In  each  case,  the  input 
and  output  are  denoted  by  subscripts  1  and  2.  In  (a)  and  (fe),  the  series  capacitor  C  and 
the  shunt  inductor  L  are  so  connected  that  each  gives  an  output  proportional  to  the 
time  derivative  of  the  input; 

Fig.  5 (a)     -r-^  C  =  I2;         (6)     -37  L  =  E$ 


(2) 


The  instantaneous  voltage  and  current  are  here  denoted  E  and  /. 
C  ,  L 


(a) 

FIG.  5.    Basic  Differentiating  Net- 
works 


(a) 


FIG.  6. 


Basic  Integrating  Net- 
works 


If  instead  the  amplitudes  of  the  frequency  components  are  denoted  E  and  I,  the  corre- 
sponding relations  for  any  particular  frequency  may  be  written: 

Fig.  5  (a)     EijuC  =  J2;         (&)     lii<»L  =  E2  (3) 


Here  j  is  the  quadrature  factor  V—  1  and  &>  =  2ir/  is  the  radian  frequency.  The  two  sets 
of  equations  are  alike  in  form  except  that  d/dt  is  replaced  by  j'w;  this  explains  why  jw  is 
sometimes  called  a  differential  operator.  It  includes  the  inseparable  two  essentials  of 
differentiation,  namely,  an  amplitude  ratio  directly  proportional  to  frequency  and  a  lead- 
ing phase  shift  of  one  quadrant. 

An  example  of  differentiation  is  the  conversion  of  a  unit  step  to  a  unit  impulse.  The 
former  has  frequency  components  of  amplitude  inversely  proportional  to  frequency, 
while  differentiation  changes  it  to  the  impulse  having  frequency  components  of  uniform. 
amplitude. 

The  two  basic  integrating  networks  are  shown  in  Fig.  6.  In  (a)  and  (6),  the  series 
inductor  L  and  the  shunt  capacitor  C  are  so  connected  that  each  gives  an  output  propor- 
tional to  the  time  integral  of  the  input  : 


~ 


(4) 


Changing  the  significance  of  voltage  and  current,  E  and  I,  from  the  instantaneous  values 
above  to  the  amplitudes  of  the  frequency  components: 


Fig.6(a)     *- 


<  JBt 


(5) 


Since    Cdt  is  replaced  by  !//«,  the  latter  is  sometimes  called  an  integral  operator.     It 


5-30 


ELECTRIC  CIRCUITS,  LINES,   AND  FIELDS 


includes  the  inseparable  two  essentials  of  integration,  namely,  an  amplitude  ratio  inversely 
proportional  to  frequency  and  a  lagging  phase  shift  of  one  quadrant.  An  example  of 
integration  is  the  conversion  of  a  unit  impulse  to  a  unit  step. 

The  basic  networks  of  Figs.  5  and  6  rely  on  simple  admittance  or  impedance  coupling, 
so  the  input  and  output  are  voltage  and  current  in  one  order  or  the  other.  Approximate 
differentiation  and  integration  can  be  obtained  by  voltage-ratio  or  current-ratio  coupling, 
the  former  being  shown  in  Figs.  7  and  8.  Each  of  these  networks  includes  a  resistor  R  in 


FIG.    7.      Voltage-ratio   Differenti- 
ating Networks 


FIG.  8. 


Voltage-ratio  Integrating 
Networks 


addition  to  a  reactor  C  or  L.  The  required  approximation  to  the  ideal  operation  in 
each  case  places,  certain  requirements  on  the  time  constant  of  the  network,  which  is 
L/R  or  OR. 

Figure  9  shows  the  meaning  of  the  time  constant  in  a  charging  or  discharging  operation 
(a)  or  (6) .  In  each  case  the  transient  has  an  exponential  variation  with  time,  and  the  time 
constant  t\  is  the  length  of  time  required  to  approach  completion  of  the  operation.  Quan- 
titatively, it  is  the  time  to  go  to  (1  -  1/e)  or  0.63  of  completion  (e  =  2.72,  the  base  of 
natural  logarithms) . 

Though  the  concept  of  charging  and  discharging  is  commonly  associated  with  energy 
storage  by  the  voltage  on  a  capacitor,  it  is  equally  applicable  to  energy  storage  by  the 
current  in  an  inductor. 

A  charging  operation  shown  in  Fig.  9  (a)  is  exemplified  by  a  voltage  step  applied  to  an 
integrating  network  of  Fig.  8.  The  integrating  operation  continues  only  for  a  duration 
less  than  the  time  constant,  so  the  time  constant  must  be  longer  than  the  required 
period  of  approximate  integration.  This  is  a  general  rule  for  such  integrating  net- 
works. 

A  discharging  operation  shown  in  Fig.  9(6)  is  exemplified  by  a  voltage  step  applied  to  a 
differentiating  network  of  Fig.  7.  The  differentiating  operation  is  prolonged  for  a  duration 
exceeding  the  time  constant,  so  the  time  constant  must  be  shorter  than  the  permissible 

duration.    This  is  a  general  rule  for 
such  differentiating  networks. 

Meeting  these  conditions  in  Figs.  7 
and  8  is  promoted  by  a  large  value  of 
series  resistance  or  a  small  value  of 
shunt  resistance,  so  the  voltage  ratio 
is  small  for  the  frequency  components 
of  major  importance. 


G.  9. 


(a)  (6) 

Tune  Constants  of  Charging  and  Discharging 


Figure  10  is  a  chart  of  the  charging  and  discharging  operations  obtained  from  the  net- 
works of  Figs.  7  and  8  in  response  to  steps  and  impulses. 

OSCILLATIONS.  If  differentiation  and  integration  are  mixed  in  a  network  by  com- 
bining CLR,  the  result  is  resonance.  The  transient  response  to  a  step  or  impulse  may  then 
be  a  damped  oscillation.  Figure  11  shows  examples  of  voltage-ratio  resonant  networks 
and  the  response  of  each  to  certain  input  transients.  A  resonant  network  is  one  having 
maximum  response  at  the  frequency  of  resonance;  an  antiresonant  network  is  one  having 
minimum  response  at  that  frequency.  Either  one  exhibits  damped  oscillations  after  a 
transient  disturbance.  The  time  constant  of  damping  in  the  series-resonant  circuit  is 
2L/R,  while  that  in  the  parallel-resonant  circuit  (with  parallel  R)  is  2CR. 

REPEATING  NETWORKS.  There  are  many  kinds  of  networks  which  respond  to  a 
step  or  impulse  with  interesting  and  significant  output  transients.  One  of  the  simplest 
is  the  integrating  network  of  Fig.  8(6)  repeated  in  successive  stages  of  a  vacuum-tube 
amplifier  (No.  524.2  in  references  7  and  8,  also  reference  38) .  Figure  12  shows  the  response 
of  n  such  networks  to  an  impulse.  The  integrating  action,  denoted  by  the  time  constant 
tit  both  delays  and  widens  the  pulse  by  virtue  of  the  energy  storage  in  each  capacitor  and 
its  subsequent  discharge  (by  repeater  action)  into  the  next  capacitor.  The  delay  of  the 
pulse  exceeds  the  widening,  so  this  system  is  a  crude  delay  network.  The  pulse  peak  is 
delayed  by  (n  —  l)£i.  For  large  values  of  n  it  is  widened  to  V2T  \/n  —  1  ti,  and  it 
approaches  the  symmetrical  shape  of  a  probability  curve. 


BEHAVIOR   OF  NETWORKS  5-31 

Input  (Ej)  Network  Output    Eg) 


(a)          D-c  on 


(5)          D-c  off 


Impulse 


D-c  on 


Integrating 


Integrating 


Integrating 


Differentiating 


Charging 


Dls'cheirging 


Discharging 


Discharging 


(e)          D-c  off  Differentiating  Discharging 

FIG.  10.    Transient  Response  of  Simple  Networks 


Input  (Ei) 


Output  ( 


A-c  off 


(C)          In 


A-c  on 


or 
Resonant 


Resonant 


Anti- resonant 


Rising 


FaHlng 


ftA  A/\  ^  ^  .> 

ri-rjTnsrw^rr 


Falling 


Falling 


T 

A-c  off  Anti-resonant  Falling 

FIG.  11.     Transient  Response  of  Oscillatory  Networks 


5-32 


ELECTRIC   CIRCUITS,   LINES,   AND  FIELDS 


The  response  of  a  resonant  circuit  to  an  impulse,  as  exemplified  in  Fig.  1 1  (c)  for  a  single 
network,  may  be  extended  to  repeating  networks.  The  envelope  of  the  resulting  transient 
oscillation  then  assumes  the  form  of  Fig.  12. 


FIG.  12.    The  Delay  and  Widening  of  a  Pulse  by  Repeating  Integrating  Networks 

BANDWIDTH.  The  integrating  action  of  a  shunt  capacitor  limits  the  speed  of  in- 
formation by  restricting  the  frequency  bandwidth.  This  is  a  major  factor  in  a  wide-band 
amplifier  for  such  uses  as  television  and  radar,  because  each  interstage  coupling  from  one 
tube  to  the  next  has  inherent  shunt  capacitance.  Figure  13  shows  a  resistance-coupled 
amplifier  stage  subject  to  inherent  shunt  capacitance  C  across  the  coupling  resistor  R. 

The  frequency  variation  of  the  response  of  such  an  amplifier  is  shown  in  Fig.  14,  in 
which  /i  is  the  frequency  at  which  the  reactance  of  the  capacitor  (l/[27r/C])  is  equal  to  the 
shunt  resistance  (R).  Figure  14(a)  shows  the  amplitude  variation  for  several  conditions, 
starting  with  (1)  simply  R  and  C.  The  bandwidth  is  increased  (2)  by  adding  inductance  L 
to  build  up  the  impedance  by  a  tendency  to  resonance.  A  more  complicated  network, 


2 

(a) 


///! 


f/A 


FIG.  13.  Amplifier  with  Shunt  Ca-  FIG.  14.  The  Limitation  of  Bandwidth  by  Shunt  Capacitance 
pacitance  Limiting  the  Bandwidth 

termed  the  "dead-end  filter"  (reference  15),  can  be  used  to  extend  the  bandwidth  as  far 
as  curve  (3)  but  no  further.  The  maximum  bandwidth  over  which  a  uniform  amplitude 
ratio  can  be  obtained  is  theoretically 


In  practice,  about  half  this  bandwidth  is  obtained  in  simple  circuits  with  a  sufficient 
approximation  to  uniformity. 

If  many  stages  of  wide-band  amplification  are  needed,  the  phase  distortion  shown  in 
Fig.  14(6)  may  be  a  limiting  factor  more  severe  than  amplitude  distortion.  The  ideal 
phase  variation  is  a  linear  proportionality  to  the  frequency.  Condition  (1),  with  simply 
R  and  C,  yields  a  convex  phase  curvature,  but  in  this  case  the  amplitude  distortion  is 
more  severe.  (See  Fig.  12,  which  applies  to  this  case.)  Condition  (2),  with  L  added, 
yields  a  concave  phape  curvature  which  happens  to  be  more  detrimental  than  the  residual 
amplitude  distortion.  There  are  cases  (notably  in  radar  pulse  receivers)  where  a  com- 
promise between  (1)  and  (2)  may  be  optimum  (reference  38).  The  extreme  condition  (3) 
yields  abrupt  changes  in  amplitude  and  phase  at  cutoff,  which  cause  transient  damped 
oscillations. 


GENERAL  PRINCIPLES  5-33 

AMPLITUDE  AND  PHASE  DISTORTION.  The  transient  response  of  linear  networks 
is  related  uniquely  with  their  steady-state  characteristics,  so  a  knowledge  of  the  latter 
makes  it  possible  to  estimate  the  transient  response.  This  is  the  province  of  the  Fourier 
integral  and  other  concepts  such  as  "paired  echoes"  for  evaluating  distortion  (references 
14  and  20).  Examples  of  amplitude  and  phase  distortion  are  shown  in  Fig.  14  for  the 
simple  case  of  Fig.  13  described  above. 

Amplitude  distortion  may  be  simply  the  limitation  of  the  bandwidth  or  may  also  include 
irregularities  within  the  bandwidth.  It  is  generally  expressed  in  terms  of  attenuation, 
since  this  is  a  logarithmic  quantity  which  can  simply  be  added  for  cumulative  stages 
(p.  1-37). 

Phase  distortion  is  any  departure  from  linear  proportionality  between  the  lagging  phase 
angle  and  the  frequency.  It  is  expressed  in  angular  measure,  which  is  additive  for  cumu- 
lative stages.  The  absolute  unit  of  angle  is  the  radian,  which  is  l/2ir  circle  or  57.3°,  The 
radian  is  quantitatively  comparable  with  the  napier,  so  6.6°  is  comparable  with  1  db. 
The  inherent  properties  of  passive  linear  networks  determine  certain  relations  between 
attenuation  and  phase  angle  (references  36  and  41) .  For  any  pattern  of  variation  of  the 
attenuation  over  the  entire  frequency  range,  there  is  a  corresponding  pattern  of  -mim'trm-m 
phase  angle  obtainable  in  networks.  Those  networks  which  provide  selective  attenuation 
with  minimum  phase  angle  are  termed  " minimum-phase"  networks  and  others  "excess- 
phase"  networks.  Minimum-phase  networks  include  self-impedance  couplings,  simple 
ladder  networks,  and  any  other  network  whose  response  can  be  expressed  as  a  product  of 
physically  realizable  self-impedance  factors.  Excess-phase  networks  include  transmission 
lines,  all-pass  phase-correcting  networks  (lattice  or  bridged-tee),  and  networks  with 
negative  mutual  inductance. 

Phase-correcting  networks  are  theoretically  possible  free  of  attenuation,  but  not  atten- 
uation-correcting networks  free  of  phase  distortion.  Therefore  it  is  customary,  in  the 
design  of  a  practical  network,  first  to  obtain  the  required  attenuation,  and  then  if  necessary 
to  add  phase-correcting  networks  for  obtaining  linear  phase.  Usually  the  phase  correction 
need  be  effective  only  over  the  band  width  of  nearly  maximum  response.  Phase  correction 
(free  of  attenuation)  always  increases  the  phase  angle,  never  the  reverse. 

Amplitude  distortion,  free  of  phase  distortion,  cannot  destroy  the  symmetry  of  a  sym- 
metrical input  pulse.  Therefore  any  asymmetrical  distortion  is  a  symptom  of  departure 
from  phase  linearity. 

A  sharp  cutoff  at  the  edge  of  the  useful  bandwidth  causes  a  damped  oscillation  or  "over- 
shoot" in  the  transient  response  to  an  impulse  or,  in  less  degree,  to  a  step.  If  this  result 
is  not  permissible,  a  gradual  cutoff  is  required,  as  shown  in  Fig.  3(c)  (reference  11). 

15.  GENERAL  PRINCIPLES 

THE  FOURIER  INTEGRAL.  In  the  study  and  design  of  networks  to  handle  transient 
disturbances,  the  most  powerful  concept  is  the  Fourier  integral.  It  is  an  extension  of  the 
more  familiar  Fourier  series,  which  is  restricted  to  periodic  waves  but  still  serves  as  an 
introduction  to  the  integral.  Each  is  essentially  a  relationship  between  a  disturbance 
over  a  period  of  time  and  its  frequency  components;  or,  conversely,  a  set  of  components 
can  be  synthesized  into  the  form  of  disturbance. 

The  following  presentation  of  the  Fourier  series  is  in  a  form  well  adapted  for  extension 
to  the  integral  and  useful  for  direct  application. 

A  wave  form  (of  voltage  or  current,  for  example)  is  denoted  T(t}  and  is  completed  in 
the  time  interval  between  —  ti/2  and  -Hi/2.  The  same  wave  form  is  repeated  hi  successive 
intervals  of  the  same  period  fr,  as  a  periodic  wave.  This  wave  can  be  expressed  as  a  sum 
of  sine-wave  components  of  harmonic  frequencies  and  the  proper  phase: 

00 

T(t)  =     S    Fn  exp  (j2*nfiQ  (7) 

n=  —  oo 

The  fundamental  frequency  is  /i  =  l/fc.  Each  harmonic  component  has  a  frequency  nfi 
which  is  an  integral  multiple  of  the  fundamental  frequency.  Its  amplitude  is  Fn,  which 
is  generally  complex  to  include  the  phase  angle.  Obviously  the  dimensional  units  of  T(f) 
and  Fn  are  the  same. 

The  sine-wave  nature  of  each  component  is  indicated  by  the  unit  vector  rotating  at  a 
frequency  nfi : 

exp  (fiirnfit)  =  cis  (2irnfit}  =  cos  (2™/i«)  -f  /  sin  (2-irnfit)  (8) 

Since  T(f)  is  real,  it  is  apparent  that  the  imaginary  parts  of  each  component  in  the  summa- 
tion must  cancel  out.  This  cancellation  occurs  between  the  amplitudes  Fn  of  the  com- 


5-34  ELECTEIC  CIRCUITS,   LINES,   AND   FIELDS 


= r 

j  -K 


ponents  of  equal  positive  and  negative  values  of  n.    The  zero-frequency  (direct)  com- 
ponent (n  =*  0)  always  has  a  real  amplitude  FQ. 

The  (complex)  amplitude  of  each  component  is  formulated  as 

"h/2 

T(t]  exp  (-ftirnfit]  dt  (9) 

'-'1/2 

This  integral  selects  and  evaluates  each  harmonic  of  frequency  nfi.    A  tabular  or  graphical 
or  formal  integration  can  be  used  to  compute  Fn,  which  in  general  will  be  complex. 

The  extension  of  the  series  to  the  integral  requires  two  more  concepts.  First,  since  a 
transient  is  presumed  to  occur  only  once,  the  period  ti  is  made  very  large  and  the  corre- 
sponding fundamental  frequency  /i  very  small.  Secondly,  the  harmonic  amplitude  Fn 
is  changed  to  the  frequency  spectrum  F(f)  which  represents  the  "amplitude-frequency 
density"  or  "amplitude  per  unit  of  frequency"  in  the  vicinity  of  the  frequency/  (instead 
of  nfi).  The  density  does  not  have  the  same  dimensional  units  as  TOO  but  rather  the 
same  units  multiplied  by  "time."  For  example,  if  T(£)  is  in  volts,  F(f)  is  in  volts  per  cycle 
per  second,  or  volt-seconds. 

With  these  steps,  there  follows  the  Fourier  integral  for  expressing  the  wave  form  of  a 
transient  in  terms  of  its  frequency  spectrum  F(f) : 

T(t)  =    C  F(f)  exp  (/27T/0  df  (10) 

J   —  00 

The  other  form  expresses  the  frequency  spectrum  in  terms  of  the  transient: 

F(f)  =    r™   T(t]  exp  (-/27T/0  dt  (11) 

J  —  00 

Following  Campbell  and  Foster  (references  7  and  8),  the  above  forms  are  symmetrical 
in  that  F(f)  and  T(t}  are  interchangeable  merely  by  reversing  the  sign  of/.  This  is  accom- 
plished by  integrating  from  —  =o  to  +  °° ,  and  expressing  in  terms  of  the  variables  /  (instead 
of  to  =  2x/)  and  t,  which  are  mutually  reciprocal.  These  forms  include  j2ir  under  the 
exponential  function,  which  is  logical  since  J%TT  is  the  natural  logarithm  of  a  unit  vector 
rotated  by  one  cycle,  and  exp  (/27r)  symbolizes  that  vector. 

In  electrical  networks,  the  interchangeability  of  F(f)  and  T(t}  has  reality  only  if  F  is 
real,  since  T  is  real.  This  is  true  only  in  idealized  networks,  which  may  be  instructive 
examples.  In  such  cases,  both  F  and  T  are  symmetrical  in  form. 

The  direct  significance  of  F(f)  and  T(t)  is  simple.  It  is  based  on  the  unique  fact  that 
an  impulse  has  a  uniform  amplitude  density  over  the  spectrum.  If  a  unit  impulse  is 
applied  to  a  network  which  modifies  its  frequency  spectrum  to  the  form  F(f),  the  output 
is  a  transient  of  the  form  T(t). 

If  a  transient  TI  of  any  shape  is  applied  to  any  network,  the  output  transient  T  can 
be  expressed  and  often  evaluated  simply  by  the  following  procedure:  (1)  compute  FI,  the 
spectrum  of  the  input  transient  TI\  (2)  formulate  F2,  the  frequency  response  of  the  net- 
work; (3)  note  that  the  frequency  spectrum  of  the  output  transient  is  F  —  FiF%',  (4)  from 
the  knowledge  of  F,  use  the  Fourier  integral  to  express  the  out- 
put transient  T.  If  the  input  is  a  unit  step  instead  of  a  unit 
impulse,  Fi  =  l//2?r/. 

Closely  related  methods  of  transient  analysis  and  synthesis  are 
the  Laplace  transformation  and  the  Heaviside  operational  calculus. 
The  Fourier  integral  is  a  restricted  case  of  the  Laplace  transform, 
but  actually  the  one  which  is  most  simply  adapted  to  the  study 
of  transients  in  electrical  networks.  The  operational  calculus  is 
merely  a  process  for  deriving,  explaining,  and  applying  some  of 
the  ideas  inherent  in  the  Fourier  integral.  The  present  state  of 
the  literature  places  the  Fourier  integral  in  a  position  to  be  of  the 
greatest  aid  in  solving  network  problems. 

THE  SUPERPOSITION  THEOREM.  Any  transient  can  be 
regarded  as  built  up  of  a  number  of  steps  or  pulses,  a  large  num- 
ber for  smooth  wave  forms.  This  is  the  basis  of  the  superposition 
theorem.  A  simple  example,  and  probably  the  first  to  be  dis- 
covered, is  shown  in  Fig.  15.  A  square  pulse  (a)  of  any  width  is 
the  resultant  of  two  superimposed  steps  (6) ,  the  first  one  positive 

_,  and  the  second  negative.    The  transient  response  of  a  network  to 

IG*  Superpositfon  *ke  sQuare  pulse  is  obtained  by  first  evaluating  its  response  to  each 

step  (c)  and  superimposing  these  transients  (d). 

The  FiFz  procedure  described  above  has  an  alternative  in  the  superposition  theorem 
of  the  operational  calculus.  This  theorem  is  encompassed  in  a  single  Fourier  integral  (No. 
202  in  references  7  and  8).  As  above,  TI  is  the  input  transient  (whose  frequency  spectrum 


GENERAL  PRINCIPLES 


5-35 


is  F{)  and  F2  is  the  response  of  the  network;  T%  is  the  transient  which  would  be  obtained 
from  the  network  in  response  to  a  unit  impulse.  The  output  transient  is  then  expressed 
in  terms  of  T\  and  Tz  as  follows: 


T(t)  =    f 

J  — 


exp 


df 


-  £')  dtf  «    f  °°   Ti(t  -  t'} 

J  —  00 


dtf 


(12) 


(13) 


The  symbol  tf  denotes  the  variable  of  integration,  as  distinct  from  i,  the  time  variable  in 
the  transient. 

THE  ENERGY  INTEGRAL.     One  of  the  most  useful  corollaries  of  the  Fourier  integral 
is  the  "energy  integral": 

r  \p\*df-  r  T*dt  (u) 

J  —  00  */-  CO 

It  states  that  the  energy  of  the  transient  is  proportional  to  the  area  under  the  energy- 
distribution  curve  over  the  frequency  range,  this  curve  being  plotted  in  terms  of  1  F  ]2. 
This  concept  has  the  widest  use  for  the  evaluation  of  the  power  associated  with  random 
noise,  which  behaves  as  a  great  number  of  impulses  occurring  at  random.  It  is  noted 
that  the  total  energy  is  determined  by  the  amplitude  (squared)  independent  of  phase. 

IDEALIZED  FILTERS.     Idealized  examples  of  niters  and  transients  are  instructive 
as  to  the  basic  limitations.    The  most  common  such  example  is  shown  in  Fig.  16  (references 


(6) 


(o) 

FIG.  16.    An  Idealized  Filter  and  Its  Transient  Response 

9  and  10).  A  network  (a)  has  uniform  response  (with  zero  phase  angle)  up  to  a  cutoff 
frequency  /c,  and  no  response  above  this  frequency.  If  an  impulse  is  applied  to  this 
network,  the  output  transient  (&)  is  symmetrical  and  is  accompanied  by  transient  oscilla- 
tions at  the  cutoff  frequency.  If  this  response  is  approximated  in  a  real  network,  the 
phase  slope  is  rather  great  and  delays  the  main  pulse  so  much  that  the  earliest  perceptible 
oscillations  (preceding  the  pulse)  occur  at  a  time  later  than  the  applied  impulse.  Inci- 
dentally, this  requires  an  "excess-phase"  network  as  defined  above. 

In  the  transient  of  Fig.  16(&),  the  nominal  pulse  duration  may  be  denned  as  2tc,  deter- 
mined by  the  dotted  rectangle  of  equal  net  area.  If  a  step  (instead  of  an  impulse)  is 
applied  to  the  same  network,  the  output  (c)  has  a  slope  of  the  same  nominal  duration 
(called  the  "slope  time"  or  "time  of  rise'*  or  "build-up  time"). 

The  ideal  filter  of  Fig.  16 (a)  has  a  perfectly  sharp  cutoff,  which  causes  transient  oscilla- 
tions in  the  output  (6)  and  (c).  The  opposite  extreme  is  a  response  having  gradual  cutoff 
of  the  form  of  a  probability  curve,  somewhat  similar  to  Fig.  3(c)  (reference  11).  In  re- 


5-36  ELECTRIC   CIRCUITS,   LINES,   AND   FIELDS 

sponse  to  an  impulse,  such  a  filter  yields  a  rounded  symmetrical  pulse  of  the  same  shape, 
free  of  overshoot  and  transient  oscillations.  This  performance  is  approximated  by  a 
large  number  of  networks  like  Fig.  8,  with  output  transients  shown  in  Fig.  12.  This  case 
has  been  found  ideal  in  radar  receivers,  to  assure  the  prompt  damping  of  one  echo  pulse 
in  readiness  for  another. 

BELAY.  A  lagging  phase  angle  is  characteristic  of  filters  passing  a  limited  band  width 
(low-pass  and  band-pass),  and  also  of  transmission  lines.  The  result  is  a  delay  of  the 
signal,  and  perhaps  also  a  distortion  of  its  wave  form. 

The  delay  is  defined  in  different  ways,  as  a  function  of  frequency.  If  /3  is  the  lagging 
phase  angle  (in  radians)  at  a  frequency  o>  (in  radians  per  second) ,  the  "  intercept  delay" 
or  "phase  delay"  (in  seconds)  is  merely  /3/co.  It  signifies  the  delay  of  each  sine-wave 
component  along  the  time  axis.  If  the  delay  is  the  same  for  all  components,  it  is  free  of 
distortion.  This  result  requires  *' linear  phase,"  that  is,  a  phase  angle  directly  proportional 
to  the  frequency. 

The  more  general  concept  is  the  "envelope  delay,"  defined  as  the  "phase  slope" 
dfi/du.  This  definition  is  free  of  the  ambiguity  of  multiples  of  2?r  in  determining  the  phase 
angle.  It  derives  its  name  from  its  significance  as  the  delay  of  the  envelope  of  a  transient 
oscillation  of  many  cycles,  as  distinguished  from  the  delay  of  the  individual  cycles.  The 
delay  of  the  cycles  is  inconsequential  if  the  transient  is  to  be  rectified,  in  which  event  the 
envelope  delay  uniquely  determines  the  effect  of  the  phase  angle  on  the  output  of  the 
rectifier. 

These  two  definitions  of  delay  correspond  with  those  of  the  wave  velocity  in  a  trans- 
mission line  or  other  wave  medium.  The  intercept  delay  or  phase  delay  determines  the 
"phase  velocity"  or  "steady-state  velocity,"  with  its  ambiguities.  The  envelope  delay 
uniquely  determines  the  "group  velocity," -which,  from  one  point  to  another,  cannot 
exceed  the  speed  of  light. 

BIBLIOGRAPHY 

1.  C.  P.  Stemmetz,  Transient  Electric  Phenomena  and  Oscillations,  third  ed.    McGraw-Hill  (1920). 

(Many  original  concepts,  mainly  superseded  by  later  methods  of  presentation.) 

2.  R.  V.  L.  Hartley,  Relations  of  Carrier  and  Sidebands  in  Radio  Transmission,  Proc.  I.R.E.,  Vol.  11, 

34-56  (February  1923).    (Double  or  single  sideband,  with  or  without  carrier.) 

3.  L.  A.  Hazeltine,  Electrical  Engineering.     The  Macmillan  Co.    (1924).     Chapter  6,  "Transient 

Currents  and  Electric  Waves,"  pp.  210-254.    (The  transient  response  in  simple  circuits  and  lines.) 

4.  H.  Nyquist,  Certain  Factors  Affecting  Telegraph  Speed,  B.S.T.J.,  Vol.  3,  324-352  (April  1924). 

(The  limitation  imposed  by  the  frequency  band  width  and  the  choice  of  a  code.) 

5.  H.  Nyquist,  Certain  Topics  in  Telegraph  Transmission  Theory,  Trans.  A.I.E.E,,  Vol.  47,  617-644 

(April  1928),  (The  relation  between  speed  and  frequency  bandwidth;  the  case  of  single  sideband 
with  half-amplitude  carrier.) 

6.  R.  V.  L.  Hartley,  Transmission  of  Information,  B,S.T.J.,  Vol.  7,  535-563  (July  1928).     (The 

amount  of  information  proportional  to  the  product  of  the  frequency  bandwidth  and  the  time.) 

7.  G.  A.  Campbell,  Practical  Application  of  the  Fourier  Integral,  B.S.T.J.,  Vol.  7,  639-707  (October 

1928).     (Purely  mathematical  introduction;  table  of  paired  coefficients.) 

8.  G.  A.  Campbell  and  R.  M,  Foster,  Fourier  Integrals  for  Practical  Applications,  Bell  Tel.  Sys.  Tech. 

PubL,  Monograph  B-584,  September  1931.  (Purely  mathematical  introduction;  the  most 
extensive  tables  of  paired  coefficients.) 

9.  A.  T.  Starr,  Transients  in  Networks,  Electric  Circuits  and  Wave  Filters,  Sir  Isaac  Pitman  &  Sons, 

Ltd.,  Chapter  12,  pp.  332-353,  1934.  (Introduction  to  the  concept  of  idealized  networks  and 
their  transient  response.) 

10.  E.  A.  Guillemin,   Communication  Networks,  Vol.  II.     John  Wiley   (1935).     Chapter  11,  "The 

Transient  Behavior  of  Filters,"  pp.  461-507. 

11.  H.  A.  Wheeler  and  A.  V.  Loughren,  The  Fine  Structure  of  Television  Images,  Proc.  I.R.E.,  Vol.  26, 

540-575  (May  1938).  (The  band-width  limitation,  especially  the  effect  of  sharp  cutoff  or  gradual 
cutoff;  a  simple  introduction  to  the  Fourier  integral  in  the  absence  of  phase  distortion;  bibli- 
ography.) 

12.  J.  E.  Smith,  B.  Trevor,  and  P.  S.  Carter,  Selective  Sideband  vs.  Double  Sideband  Transmission  of 

Telegraph  and  Facsimile  Signals,  R.C.A.  Rev.,  Vol.  3,  213-238  (October  1938). 

13.  A.  V.  Bedford  and  G.  L.  Fredendall,  Transient  Response  of  Multistage  Video-frequency  Amplifiers, 

Proc.  I.R.E.,  Vol.  27,  277-284  (April  1939). 

14.  H.  A.  Wheeler,  The  Interpretation  of  Amplitude  and  Phase  Distortion  in  Terms  of  Paired  Echoes, 

Proc,.  I.R.E.,  Vol.  27,  359-385  (June  1939), 

15.  H.  A.  Wheeler,  Wide-band  Amplifiers  for  Television,  Proc.  I.R.E.,  Vol.  27,  429-438  (July  1939). 

(Fundamental  limitations  imposed  by  shunt  capacitance;  idealized  networks  including  dead-end 
niters.) 

16.  R.  TJrtel,  Observations  Regarding  Television  Transmission  by  Single  Sideband,  Telefunken  Haus- 

mitteilungen,  No.  81,  pp.  80-83  (July  1939).  (The  envelope  distortion  in  the  transmission  of 
single  sideband  and  half-amplitude  carrier  through  a  filter  of  idealized  characteristics.) 

17.  Leon  S.  Nergaard,  A  Theoretical  Analysis  of  Single-sideband  Operation  of  Television  Transmitters, 

Proc.  I.R.E.,  Vol.  27,  666-677  (October  1939). 

18.  S.  Goldman,  Television  Detail  and  Selective-sideband  Transmission,  Proc.  I.R.E.,  Vol.  27,  725-732 

(November  1939).  (The  envelope  distortion  in  the  transmission  of  single  sideband  and  half- 
amplitude  carrier  through  a  filter  of  idealized  characteristics.) 

19.  H.  Nyquist  and  K.  W.  Pfleger,  Effect  of  the  Quadrature  Component  in  Single-Sideband  Trans- 

mission, B.S.T.J.,  Vol.  19,  63-73  (January  1940).  (The  envelope  distortion  in  the  transmission 
of  a  single  sideband  and  half-amplitude  carrier  through  a  filter  of  idealized  characteristics^ 


NON-LINEAR   ELECTRIC    CIRCUITS  5-37 

20.  Von  F.  Strecker,  The  Influence  of  Small  Phase  Distortion  on  the  Reproduction  of  Television 

Signals,  E.N.T.,  Vol.  17,  51-56  (March  1940);  The  Effect  of  Amplitude  and  Phase  Distortion, 
93-107  (May  1940).  (Paired  echoes.) 

21.  R.  D.  Kell  and  G.  L.  Fredendall,  Selective  Sideband  Transmission  in  Television,  R.C.A.  Rev.,  Vol.  4, 

425-440  (April  1940) .    (Response  of  single-sideband  receiver  to  a  unit  step  of  modulation.) 

22.  M.  I.  T.  Engineering  Staff,  Electric  Circuits.    John  Wiley  (1940).     (A  comprehensive  treatment 

of  networks  with  emphasis  on  their  transient  response.) 

23.  H.  E.  Kallmann,  Transversal  Filters,  Proc.  I.R.E.,  Vol.  28,  302-310  (July  1940).     (The  synthesis, 

by  a  pattern  of  echoes,  of  the  signal  which  would  be  produced  by  an  idealized  filter  free  of  phase 
distortion.) 

24.  H.  E.  Kallmann  and  R.  E.  Spencer,  Transient  Response  of  Single-sideband  Systems,  Proc.  I.R.E., 

Vol.  28,  557-561  (December  1940).  (The  envelope  distortion  in  the  transmission  of  single  side- 
band and  half -amplitude  carrier  through  a  filter  of  idealized  characteristics.) 

25.  C.  P.  Singer,   A  Mathematical  Appendix  to  Transient  Response  of  Single-sideband  Systems, 

Proc.  I.R.E.,  Vol.  28,  561-563  (December  1940). 

26.  H.  S.  Carslaw  and  J.  C.  Jaeger,  Operational  Methods  in  Applied  Mathematics.    Oxford  University 

Press  (1941).    (The  Laplace  transformation.) 

27.  H.  A.  Wheeler,  The  Solution  of  Unsymmetrical-sideband  Problems  with  the  Aid  of  the  Zero- 

frequency  Carrier,  Proc.  I.R.E.,  Vol.  29,  446-458  (August  1941). 

28.  H.  A.  Wheeler,  Common-channel  Interference  between  Two  Frequency-modulated  Signals,  Proc. 

I.R.E.,  Vol.  30,  34-50  (January  1942).  (The  response  to  frequency  modulation,  by  an  analysis 
which  is  general  for  any  wave  form  of  modulation,  either  transient  or  periodic.) 

29.  J.  B.  Russell,  Heaviside's  Direct  Operational  Calculus,  Elec.  Eng.,  Vol.  61,  84-88  (February  1942). 

(A  brief  introduction.) 

30.  R.  V.  L.  Hartley,  A  More  Symmetrical  Fourier  Analysis  Applied  to  Transmission  Problems,  Proc. 

I.R.E.,  Vol.  30,  144-150  (March  1942). 

31.  W.  L.  Sullivan,  Fourier  Integrals,  Elec.  Eng.,  Vol.  61,  248-256  (May  1942).     (An  excellent  intro- 

duction and  bibliography.) 

32.  H.  Salinger,  Transients  in  Frequency  Modulation,  Proc.  I.R.E.,  Vol.  30,  378-383  (August  1942.) 

33.  A.  V.  Bedford  and  G.  L.  Fredendall,  Analysis,  Synthesis,  and  Evaluation  of  the  Transient  Response 

of  Television  Apparatus,  Proc.  I.R.E.,  Vol.  30,  440-457  (October  1942). 

34.  M.  F.  Gardner  and  J.  L.  Barnes,  Transients  in  Linear  Systems,  Vol.  1,  "Lumped-constant  Systems." 

John  Wiley  (1942).  (The  most  intensive  treatment  of  the  subject,  based  on  the  Laplace  trans- 
formation, with  emphasis  on  electromechanical  analogues.  A  complete  bibliography.) 

35.  J.  G.  Brainerd,  Ultra-High-Frequency  Techniques.    D.  Van  Nostrand  (1942).    Chapter  1,  "Linear 

Circuit  Analysis,"  pp.  1-47.    (Transient  response  to  steps  and  pulses.) 

36.  F.  E.  Terman,  Relation  between  Attenuation  and  Phase  Shift  in  Four-terminal  Networks,  Proc. 

I.R.E.,  Vol.  31,  233-240  (May  1943). 

37.  Harold  Pender  and  S.  R.  Warren,  Electric  Circuits  and  Fields.    McGraw-Hill  (1943).    Chapter  4, 

"Introduction  to  Transient  Phenomena  in  Electric  Circuits,"  pp.  108-133. 

38.  W.  W.  Hansen,  Transient  Response  of  Wide-band  Amplifiers,  Proc.  of  National  Electronics  Confer- 

ence, Vol.  1,  544-554  (1945);  also  Electronic  Industries,  Vol.  Ill,  No.  11,  80-82,  218,  220 
(November  1944).  (A  definition  of  bandwidth  for  transients,  and  its  application  to  amplifiers 
of  one  to  100  stages.) 

39.  R.  V.  Churchill,  Modern  Operational  Mathematics  in  Engineering.     McGraw-Hill  (1944). 

40.  W.  W.  Hansen,  On  Maximum  Gain-bandwidth  Product  in  Amplifiers,  J.  Applied  Phys.,  Vol.  16, 

528-534  (September  1945). 

41.  H.  W.  Bode,  Network  Analysis  and  Feedback  Amplifier  Design.  _  D.  Van  Nostrand  (1945).    (A  com- 

prehensive treatment  with  emphasis  on  stability  and  bandwidth.) 

42.  H.  E.  Kallmann,  R.  E.  Spencer,  and  C.  P.  Singer,  Transient  Response,  Proc.  I.R.E.,  Vol.  33, 

169-195  (March  1945). 

43.  M.  Levy,  The  Impulse  Response  of  Electrical  Networks,  Elec.  Com.,  Vol.  22,  No.  1,  40-55  (1944). 

(Idealized  networks.) 

44.  G.  W.  Carter,  The  Simple  Calculation  of  Electrical  Transients.    The  Macmillan  Co.  (1945).     (A 

short  mathematical  course  based  on  operational  methods,  with  well-selected  tables  and  bibli- 
ography.) 

45.  Ernest  Frank,  Pulsed  Linear  Networks.    McGraw-Hill  (1945).     (An  introductory  course  limited 

to  the  direct  application  of  differential  equations.) 

46.  D.  G.  Tucker,  Transient  Response  of  Filters,  Wireless  Engineer,  Vol.  XXIII,  No.  269,  pp.  36-42 

(February  1946);  VoL  XXIII,  No.  270,  84-90  (March  1946). 

Note:  Reference  31  is  recommended  as  a  good  introduction  to  the  more  advanced  concepts,  and  a 
good  bibliography  to  that  date.  References  29  and  31,  with  three  related  papers,  have  been  reprinted 
by  A.I.E.E.  under  the  title  Advanced  Methods  of  Mathematical  Analysis  (1942). 


NON-LINEAR  ELECTRIC  CIRCUITS 

By  Knox  McHwain 
In  solving  the  fundamental  electric  circuit  equation 


the  assumption  is  often  made  that  r,  L,  and  C  are  constants.  Circuits  in  which  these 
conditions  exist  are  called  "linear  circuits"  and  have  been  treated  above.  If  any  one  or 
more  of  these  circuit  parameters  vary  with  the  current  through  it  or  the  voltage  across  it, 
the  solution  of  eq.  (1)  hi  general  takes  the  form  of  an  infinite  series^  so  that  the  current 
wave  form  is  not  a  replica  of  the  voltage  wave  form.  Such  circuits  are  called  "non- 
linear circuits." 


5-38  ELECTRIC   CIRCUITS,  LINES,   AND  FIELDS 

NON-LINEAR  DISTORTION.  When  it  is  desired  that  the  wave  form  of  the  current 
through  a  particular  circuit  element  be  the  same  as  the  wave  form  of  the  original  voltage, 
and  when  the  response  of  the  element  is  not  directly  proportional  to  the  driving  force 
(non-linear  circuit),  the  element  is  said  to  introduce  non-linear  distortion.  When  the 
input  voltage  is  a  simple  sine  wave,  the  output  current  will  contain  components  of  double 
and  higher  multiples  of  the  impressed  frequency,  the  amplitudes  being  determined  by 
the  series  solution  mentioned  above. 

Non-linear  distortion  is  most  serious  in  sound  reproduction  since  the  spurious  sound 
harmonics  are  readily  noted  by  the  ear  and  produce  a  very  unpleasant  sensation  for  the 
fastidious  listener.  If  the  rms  value  of  the  introduced  harmonics  is  kept  below  5  per  cent 
of  the  rms  value  of  the  fundamental,  the  non-linear  distortion  will  not  be  objectionable 
(sometimes  even  10  per  cent  is  allowed). 

One  important  characteristic  of  this  type  of  distortion  is  that,  once  introduced,  it  is 
difficult  if  not  impossible  to  correct  for  it,  so  that  the  component  parts  of  the  circuit  must 

be  separately  designed  so  as 
to  have  linear  voltage  versus 
current  characteristics  (except 

Curjeni  for  modulators  and  detectors, 

see  Section  7) . 

SOLUTION  OF  NON- 
LINEAR CIRCUITS.  The 
change  in  inductance  due  to 
the  non-linear  B-H  curve  of 
iron,  the  varying  resistance 
of  an  electric  arc  or  of  the 
thermionic  vacuum  tube,  or 
the  varying  capacitance  of 
the  condenser  microphone,  all 
introduce  non-linear  effects. 
The  simplest  case  is  where  the 
resistance  is  some  function  of 
Voltage  voltage  or  current.  Fre- 

FIG.  1.    Non-linear  Current-voltage  Characteristic  quently  this  functional  rela- 

tionship is  given  as  a  curve  of 

current  against  voltage,  as  in  Fig.  1.    The  first  step  is  to  fit  an  analytic  expression  to  the 
curve.    Two  methods  have  been  widely  used  in  communication  practice: 

(a)  a  power  series  of  the  form 

i  -  A0  +  Ai  e  +  A2  e*  -f  A,  e*  +  •  -  -  (2) 

(6)  a  trigonometric  series 

i  -  AQ  +  Ai  sin  (e  +  ft)  +  A*  sin  (2«  +  0a)  +  •  *  •  (3) 

16.  POWER  SERIES  SOLUTION 

Such  devices  as  thermionic  vacuum  tubes,  electric  arcs,  some  surface  contacts  (such  as 
copper  oxide  and  lead),  and  certain  crystalline  structures  have  a  current-voltage  char- 
acteristic similar  to  that  shown  in  Fig.  1.  Since  the  instantaneous  value  of  the  electric 
resistance  to  an  increment  of  current  is  Ae/Ai,  the  resistance  varies  with  the  applied  volt- 
age, or  r  —  F(e). 

The  coefficients  of  the  power  series  of  eq.  (2)  may  be  evaluated  as  follows:  If  four 
terms  of  the  power  series  are  of  interest,  choose  four  separated  points  on  the  current- 
voltage  curve  and  use  the  four  sets  of  values  of  e  and  i  in  eq.  (2) .  Solve  the  four  resulting 
algebraic  equations  by  any  of  the  standard  methods  for  AI,  A%,  etc.  This  method  may 
always  be  used  no  matter  how  irregular  the  curve  and  will  always  give  correct  results 
for  the  points  chosen;  unless  the  curve  is  smooth  it  may  give  quite  incorrect  results  for 
intermediate  points.  The  characteristic  must  be  plotted  including  all  resistance  in  the 
circuit  whether  variable  or  invariable.  It  is  extremely  difficult  to  include  the  effect  of 
reactance  in  the  circuit.  The  application  of  the  method  is  thus  limited,  but  it  is  useful 
in  certain  special  cases. 

TAYLOR'S  SERIES.  If  the  equation  of  the  current-voltage  characteristic  is  known, 
or  if  it  can  be  found  (it  is  frequently  of  the  form  i  —  Ke^  for  at  least  portions  of  the  curve, 
in  which  case  the  constants  can  be  evaluated  by  plotting  on  logarithmic  cross-section 
papfer),  the  solution  by  Taylor's  series  is  useful  in  evaluating  the  coefficients  of  eq.  (2). 


TRIGONOMETRIC  SERIES  5-39 

The  current  at  any  point  (e)  in  terms  of  the  current  at  a  particular  point  (EQ)  can  be 
written. 

*- 


where  the  derivatives  are  all  evaluated  at  the  operating  point  (Eo).  Care  must  be  exercised 
in  using  this  formula  that  the  voltage  is  confined  to  the  region  wherein  the  curve  follows 
the  assumed  law  and  within  the  limits  of  convergence  of  the  series.  A  detailed  treatment 
of  this  method  is  given  in  article  20. 

17.  TRIGONOMETRIC  SERIES 

In  some  applications  of  devices  such  as  vacuum  tubes  and  gas-filled  tubes,  the  devices 
operate  over  a  range  large  compared  with  the  part  of  the  characteristic  shown  in  Fig.  1. 
The  application  of  a  power  series  would  usually,  under  such  conditions,  require  an  un- 
reasonable number  of  terms  in  order  to  obtain  even  a  fair  approximation.  On  the  other 
hand,  a  trigonometric  series  with  a  properly  chosen  fundamental  period  permits  the  ex- 
pansion of  such  a  curve  using  only  a  few 
terms  for  a  fair  approximation. 

The  trigonometric  series  is  periodic  by 
nature   and  so   does  not   represent  the 


\ 


-B 


characteristic  outside  of  the  interval  over     A \ — 

which  the  harmonic  analysis  was  taken.  \ 

When  the  curve  is  of  the  form  dbc  in  Fig.  2     ot ^ 

and  the  voltage  varies  only  over  the  in- 
terval dbe,  the  curve  can  be  arbitrarily     FIG.  2.    Construction  for  Use  of  Trigonometric  Series 
replaced  by  any  other  curve  outside  of 

this  interval.  If  the  curve  outside  of  this  interval  is  replaced  by  a  curve  as  shown  by  the 
broken  line  in  Fig.  2,  such  that  the  resulting  curve  is  skew-symmetric  about  the  point  e  and 
symmetrical  about  the  vertical  axis  through  d,  eq.  (3)  reduces  to 

i  »  j!0  4-  AI  sin  e  +  AI  sin  3e  +  •  •  •  (5) 

Because  of  the  symmetrical  properties  of  the  resulting  curve  both  the  even  harmonics 
and  the  phase  angles  become  zero.  The  choice  of  curve  outside  of  the  operating  interval 
so  that  the  resulting  curve  has  these  properties  of  symmetry  is  always  possible.  The 
convergence  of  the  series  is  usually  greatly  increased  by  using  this  built-up  curve  (with 
a  fundamental  period  four  times  the  distance  from  d  to  0).  However,  other  choices  of  the 
curve  outside  of  the  working  interval  are  even  better  in  particular  cases. 

A  fundamental  period  having  been  chosen  and  the  curve  having  been  completed  through- 
out this  period,  the  coefficients  of  the  harmonic  sine  functions  can  be  determined  by  means 
of  one  of  the  several  available  schemes  for  harmonic  analysis  (see  article  24). 

In  most  electrical  engineering  the  e  has  the  form  of  a  constant  voltage  plus  a  series  of 
sine-wave  voltages.  When  voltages  of  this  form  are  substituted  in  eq.  (5)  and  the  suc- 
cessive terms  expanded  by  means  of  the  trigonometric  formulas  for  the  sine  and  cosine  of 
the  sum  of  two  angles,  the  series  consists  of  terms ^of  the  form  cos  (x  sin  d)  and  sin  (a;  sin  d) 
which  can  be  expanded  by  the  formulas  of  Jacobi 

eo 

cos  (x  sin  d)  =  Jo(x)  +  2  2  Ju(x)  cos  2Rd 
and 

oo 

sin  (x  sin  d}  =  2%J  25-1  (a?)  sin  (222  -  l)d 

where  J»(a?)  is  a  Bessel  function  of  the  first  kind,  of  order  n  and  modulus  x.  These  functions 
can  be  found  in  tables  of  Bessel  functions.  See  p.  1-37. 

By  means  of  this  expansion  the  amplitudes  of  the  sinusoidal  components  of  the  current 
flowing  in  any  circuit  can  be  calculated  from  the  current-voltage  characteristic  of  the 
circuit  provided  that  the  circuit  contains  no  reactive  elements.  No  general  analytic 
method  has  yet  been  devised  for  the  direct  application  of  the  trigonometric  series  in  a 
circuit  containing  reactive  impedance  analogous  to  that  treated  in  article  20.  However, 
eq.  (3)  or  (5)  having  been  obtained,  the  differential  coefficients  of  eq.  (4)  can  be  obtained 
simply  by  differentiation. 

This  solution  can  be  extended  to  the  case  of  two  or  more  independent  variables  as  in 
the  application  to  the  three-electrode  vacuum  tube  with  variable  amplification  factor  and 


5-40  ELECTRIC   CIRCUITS,   LINES,   AND  FIELDS 

with  independent  voltages  applied  in  both  the  grid  and  plate  circuits.    However,  the  work 
is  so  voluminous  that  it  is  practicable  only  in  very  special  cases, 

18.  INDUCTANCE  VARIATION 

When  the  variation  of  inductance  is  of  importance  the  magnetization  curve  (locus  of 
the  tips  of  the  hysteresis  loops)  of  the  iron  forming  the  (closed)  magnetic  circuit  of  the 
reactor  is  required.  This  is  usually  similar  in  form  to  the  curve  in  Fig.  1  with  flux  density 
CB)  as  ordinates  and  magnetizing  force  (H)  as  abscissas.  This  curve  must  be  replotted 
in  the  form  of  i  (the  exciting  current)  as  a  function  of  <j>  (the  total  flux).  Then  by 
Taylor's  formula 


in  which  the  derivatives  are  evaluated  at  the  operating  point.    Define 

"1.1 

60  Jo       LQ 

where  LO  is  the  usual  incremental  inductance  (apparent  inductance  to  a  changein  current) 
at  the  operating  point.    Then  (see  article  20) 


and  so  forth.    Also  e  =  —  d<f>/dt  so  that    /  e  dt  =  —  <j>.    If  increments  are  considered 

<t>  —  0o  =  0a  —  —  I  ea  dt 

If  ea  =  SEn  cos  (unt  +  0n)  as  is  usual  (see  article  21) ,  then 

<t>a  -  -ea\  X  -^  I 

L     j^J 

where  the  symbol  [X]  indicates  that  the  maximum  value  of  each  first-degree  cosine  term 
of  different  frequency  in  the  e  immediately  preceding  is  to  be  multiplied  by  the  modulus 
of  the  complex  quantity  within  [X],  and  the  phase  angle  of  the  complex  (hi  this  case  ir/2) 
is  to  be  added  to  the  phase  of  the  cosine,  both  modulus  and  phase  being  evaluated  at  the 
frequency  of  the  given  cosine  term  (see  article  20) . 
Substituting  these  expressions  in  eq.  (6) 


Hence  if  the  B-H  curve  were  a  straight  line  dL/d<f>  =  0  and 

^  En  COS  (tOnt  +  &n  ~   7T/2) 


To  obtain  i  in  terms  of  the  applied  voltage  E'a  in  the  circuit  containing  the  variable 
inductance  and  a  constant  impedance  z  the  same  process  employed  in  introducing  imped- 
ance load  in  the  case  of  a  varying  resistance  must  be  employed  (see  article  21) 
There  results 

i  =  e'[  X  PI]  +  «*[  X  Qd  +  •  •  •  (8) 

where 

1         _  1 

Ci  =  ; — r~y   =  —, 

Z  +  JuLo         T 


which  is  to  be  so  interpreted  that 
and 


_  ]     £.(c,)»  f     a 

08  -     ^+  ~ 


APPROXIMATE  SERIES  EXPANSION 
which  is  to  be  so  interpreted  that 

_    clmC2(n+g)z(n+g)   +  CinCa 


5-41 


, 


19,  CAPACITANCE  VARIATION 

When  the  variation  of  capacitance  is  of  importance,  as  in  such  devices  as  the  condenser 
transmitter,  eq,  (1)  may  be  rewritten  as 

and  the  characteristic  of  the  device  plotted  as  charge  against  voltage.  The  same  methods 
of  solution  used  for  resistance  variation  are  then  available  (with  the  exception  that  the 
operator  z  introduced  in  the  detailed  solution  of  article  21  must  be  replaced  by  /wz) . 


20.  APPROXIMATE  SERIES  EXPANSION  FOR  THE  PLATE  CURRENT 
OF  A  TRIODE  (ASSUMES  |X  CONSTANT) 

Circuits  containing  thermionic  vacuum  tubes  are  chosen  to  exemplify  the  detailed 
treatment  of  non-linear  electric  circuits  because  of  the  commanding  importance  of  the 
thermionic  tube  in  communication 
practice.  This  wide  use  has  occurred 
largely  because  the  triode,  tetrode, 
etc.,  combine  an  amplification  or  con- 
trol function  with  the  non-linearity 
of  the  current-voltage  device;  in 
most  tubes  it  is  possible  to  employ 
either  or  both  functions  by  a  simple 
shift  in  electrode  operating  voltages 
(see  Section  4  for  nomenclature  and 
characteristics) . 

The  triode  is  generally  used  in 
practice  with  steady  voltages  applied 
to  both  grid  and  plate,  and  in  addi- 
tion at  least  one  varying  voltage  on 
one  of  the  control  electrodes,  and 
frequently  with  one  or  more  varying 
voltages  impressed  in  both  the  plate 
and  grid  circuits;  these  instantaneous 
currents  (it,  ic)  and  voltages  (65,  ec) 
may  be  conveniently  split  up  as 
follows: 

ib  =  It  4-  ip 
ic  =  Jo  +  v 


70 
60 
50 
40 

30 

15 

10 
9 
8 

7 

5    4 
3 

2 

1..5 

1.0 

f  { 

// 

7 

.Norrr 

a   Plate  Voltage  and 
ri  ament  Current 

f 

Ec=0 

-d 

/I 

/ 

1 
1 

1 
1 

I 

/ 

1 

/ 

/ 

1 

f 

/ 

1 

1 

\ 

~-c* 

-1 

4.  t 

1 

Expon 

ent« 

1.86 

t 

/ 

^ 

/ 

/\ 

f 

/ 

I       4      5     6   7  8  9  10         15      20         30      40    50  6C 
•^~t-Ec-EquivaIent  Total  Grid  Voltage  irt  Volts 

e&  —  jE/6  -f-  eP 

where  the  capital  letters  (Ej,,  Ec,  J&, 

7C)  indicate  steady  values  obtaining 

before  the  application  of  any  varying 

voltages  in  the  plate  or  grid  circuits; 

the  lower-case  letters  with  subscript 

p  or  g  indicate  instantaneous  values 

of  varying  components.     All  these 

voltages  are  specified  as  actual  voltages  between  electrodes;  if  the  external  circuits  contain 

impedance  these  voltages  will  differ  from  the  supply  voltages. 

The  plate  current  of  the  usual  triode  is  a  function  of  ee,  e&,  and  the  amplification  factor 


FIG.  3.    Mutual  Characteristic  of  a  Triode.    Normal  plate 
voltage  =  130  volts,  n  »  5.45. 


5-42  ELECTRIC   CIRCUITS,  LINES,   AND  FIELDS 

ju(=  —  deb'dec  when  ib  is  held  constant)  such  that  for  at  least  portions  of  the  operating 
range 

ib  =  K  (-  +  ec  Y  (10) 

where  K  and  77  are  constants.    The  values  may  be  found  for  any  tube  by  plotting  the  plate 
current  of  the  tube  against  equivalent  grid  voltage  (  --  f-  ec  J  on  logarithmic  cross-section 

paper  as  shown  in  Fig.  3.    The  value  of  77  in  most  triodes  usually  lies  between  1.5  and  2.5; 
since  it  varies  around  2.0  the  triode  is  sometimes  said  to  follow  a  square  law.    This  may 
be  sufficiently  accurate  for  rough  calculations  but  is  not  satisfactory  in  an  exact  analysis. 
Expanding  by  Taylor's  series 


where  the  subscript  0  after  a  bracket  indicates  that  the  derivatives  are  to  be  evaluated 
for  eb  —  EI,  and  ec  —  28  e.  Note  that  I&  =  F(JEt>,  Be]  and  may  be  subtracted  from  each 
side  of  eq.  (11).  Also  define  the  static  plate  resistance  as 


(12a) 
then 


1  deb  Jo 
and 

^2 


TP 
Making  these  substitutions,  eq.  (11)  becomes 

.-    _  ^  ' 


+  ^eg)  _  (ep  +  M%)2  drp~\       2(ep  +  ^g)s  /Bri\  V 
rp  2rp*         deb]o  3!rp»         \deb]J 


If,  as  is  usual  in  amplifiers,  ep  —  0,  then  the  factor  }j./rp(=  dib/dec)  frequently  appears  in 
this  equation.  It  is  designated  gm  and  called  the  grid-plate  transconductance  (mutual 
conductance).  Expansions  in  terms  of  transconductance  are  particularly  useful  when  n 
is  variable  or  when  the  plate  resistance  of  the  tube  is  so  high  that  external  resistances  are 
negligible  (see  article  22). 

21.  CHARACTERISTICS  OF  TRIODE  WITH  LOAD 

When  the  triode  is  used  with  external  impedance  there  are  voltage  drops  in  these 
impedances  so  that  the  electrode  voltages  are  the  differences  between  the  applied  voltages 
and  these  impedance  drops.  Since  eq.  (13)  applies  to  the  voltages  between  the  electrodes, 

account  must  be  taken  of  the  impedance  drops. 

RESISTANCE  LOADS.  When  the  external  loads 
are  pure  resistances  the  electrode  currents  and  voltages 
may  be  split  up  as  follows  (see  Pig.  4)  . 


FIG.  4.    Triode  with  Resistance  Load  e&  =  Eb  +  ep  =  E'b  —  rib  +  efp  —  rip 

where  ep  is  the  sum  of  the  externally  impressed  voltage  and  the  load  resistance  drop  in  the 
plate  circuit,  eg  is  similar,  and  r  and  re  are  the  external  resistances  in  the  plate  and  grid 
circuit.  The  voltages  e'g  and  e'P  are  the  driving  voltages  introduced  in  the  grid  and  plate 
circuits. 


CHARACTERISTICS   OF  TRIOBE  WITH  LOAD  5-43 

If  the  assumption  is  made  that  the  plate-grid  transconductance  gn(^  dic/deb)  is  zero, 
as  is  usual  for  small  grid  voltages,  then  the  grid  current  (rg  is  the  internal  grid  resistance)  is 


-L 

* 


and  the  varying  voltage  between  grid  and  cathode  is 

eV*      ,       eVr,rg     drg 
€*==^T^e  +  2(rg  +  re^c+'- 

The  varying  component  of  the  plate  current  (let  e  =  eg  +  efp/fj.)  is 

=      W      _       M2e%       dri\        u?e*rp(2rp  -  r) 
p       rp  +  r       2l(rp  +  r)»  ^  V       3l(r,  +  r)« 


3!(rp 


Usually  the  higher-order  derivatives  rapidly  decrease  in  value,  and,  since  in  the  higher- 
order  coefficients  the  power  of  (rp  +  r)  in  the  denominator  rises  rapidly,  all  terms  beyond 
the  third  may  be  neglected  for  most  practical  applications  of  the  tube. 

IMPEDANCE  LOADS.  When  impedances  are  connected  in  the  external  circuits  of 
the  tube  (instead  of  the  resistances  of  Fig.  4)  it  is  impossible  to  write  the  external  imped- 
ance drop  in  vector  form  until  the  plate  circuit  is  specified.  Use  may  be  made  of  the 


operational  impedance     z  =  r  +  L  —  -4-  ^J^     "whkh  is  such  that  zi  represents  the  im- 

pedance drop  regardless  of  the  form  of  i.  Of  course  it  is  impossible  to  evaluate  zi  until  i  is 
known.  The  plate  current  may  be  written  in  terms  of  an  operator  denoting  a  delayed 
multiplication,  called  the  square  cross  bracket,  as 

ip  =  e[  X  ci]  +  e2[Xcd  +  e»[Xcd+-.-  (17) 

where 

-r- 


in  which  z  is  to  be  separately  evaluated  for  each  component  frequency  of  interest.  Note 
that  z7  is  the  total  impedance  of  the  plate  circuit  at  the  particular  frequency.  Likewise 

±*ii  <s£  (19) 

2  de6J0    z7 

in  which  each  of  the  c's  is  to  be  evaluated  at  the  frequency  of  one  of  the  original  frequencies 
beating  together  and  z'  is  to  be  evaluated  at  the  beat  frequency.  For  instance,  to  find  the 
size  of  the  current  component  of  periodicity  (COOT  +  ««)  caused  by  the  beating  of  two  terms 
of  periodicity  com  and  con,  evaluate 

T?    TP  r 
EwlEnC2(m+n)   =    - 


,  ,  " 

Z  mZ  nZ  (m+n)  <Jvb  Jo 

Similarly  eg,  which  embraces  terms  caused  by  three  of  the  originally  impressed  com- 
ponents beating  together  (these  are  very  important  in  calculating  the  amount  of  interfer- 
ence introduced  in  a  radio  receiver  by  neighboring  carriers),  is 


which  is  to  be  so  interpreted  that 

n)  &>"] 

"  7^   L 
Vet  JO 


Z  <m-Kt+g) 


5-44 


ELECTRIC   CIRCUITS,   LINES,   AND   FIELDS 


Table  1.    Plate  Currents  of  a  Triode 

1.  No  load  and  resistance  load;  n  assumed  constant. 

ip  =  ae  •+•  c2e2  +  cge3  +  etc.  (e  =  eg+—2 


ci 

0 

e. 

No  load, 
aq.  (11) 

V 

-4;  21 

3^1  Vlte6_U 

^rpaVp-,    1 
2    de&2Jo/ 

Resistance 
load,  «q. 
(16) 

j" 

/i*  rs       arP-| 

si^rM**1 

r)  f  3rpl  V       i 

d'rp-,   \ 

rp+r 

2l(r,  +  r)»*J. 

Vde&Jo/ 

PrP  +  r    ^^ 

2.  Impedance  load;  ^  assumed  constant. 

ip  =  e[Xci]  +  e2[Xc2]  +  e3[Xcs]  +  etc. 


Cl 

C2 

C3 

M               M 

rpdrp-i   (Cl)2 

ci(c2z)  drp-i         rp  (ci)»  /xdr,-.  x  2       rp  a^-i  \ 

rp  +  z       z' 

eq.  (18) 

eq.  (19) 

eq.  (20) 

The  expression  e2[X  c2]  means  that  e2  is  to  be  reduced  to  first-degree  cosine  terms,  after  which  each 
cosine  term  is  to  be  multiplied  by  the  modulus  of  c2,  evaluated  at  the  frequency  or  frequencies  con- 
tained in  that  term,  and  the  phase  angle  of  c2  is  to  be  added  to  the  phase  of  the  cosine  term.  For 
the  method  of  assigning  frequencies  in  c2  and  cs  see  eqs.  (19a)  and  (20a). 

As  in  the  case  of  resistance  load  the  grid  voltage  eg  does  not  equal  the  voltage  introduced 
into  the  grid  circuit  (<3'g),  but  can  be  obtained  therefrom,  as 


-  •  -  •  (21) 

in  which  z"  is  the  total  impedance  of  the  grid  circuit.  In  the  second  term  (z")2  is  to  be 
treated  like  the  term  (c)2  in  eq.  (19),  that  is  evaluated  at  the  separate  frequencies  which 
are  beating  together. 

For  example,  assume  a  pure  cosine  voltage  e'p  —  E'ps  cos  (u8t  +  08)  impressed  in  the 
plate  circuit,  and  a  similar  voltage  eg  =  Egn  cos  (coni  +  0n)  between  grid  and  filament  (or 
anywhere  in  the  grid  circuit  if  the  grid  is  held  negative).  Then  as  far  as  Ci  and  c2  are 
concerned 

^'gn  COS  (oant  +  I 


^pa   =  : 


9V)       E'f,  cos  (ust  +  i 


/32n) 


'Z'  (2s) 

,  cos  ( 


[ 


r) 
p8  cos 


(22) 


where  * 


=  8n  -  tan"1 


•  and  0%  is  similar 


/52n  «  20V  —  tan"1    X(2n)    and  /33a  is  similar 


and 


=  0V  +  0'.  =  tan 


=  0%  —  0V  -  tan"1 


*  Note  carefully  the  positive  sign  associated  with  tan 


;  in  /35rt.     This  occurs  through  the 


subtraction  of  B'n  from  0'8,  even  though  Zn  is  in  the  denominator.    The  importance  of  this  difference 
in  sign  of  this  term  in  (S^n  and  fi$n  is  brought  out  in  Section  7,  article  18. 


METHOD   OF   SUCCESSIVE  APPROXIMATIONS  5-45 

A  comparison  of  the  expressions  for  ci  and  c2  given  by  eqs.  (18)  and  (19),  respectively, 
shows  the  effect  of  the  impedance  load,  which  appears  in  the  denominator  of  GZ  to  a  higher 
power,  in  decreasing  the  percentage  value  of  the  harmonic  terms,  or  in  straightening  the 
plate  and  mutual  characteristics.  From  this  it  may  be  seen  that  an  impedance  load  acts 
similarly  to  a  resistance  load  in  tending  to  decrease  the  relative  size  of  the  harmonics  intro- 
duced by  the  non-linear  shape  of  these  characteristics.  The  impedance  load  of  course 
introduces  frequency  distortion. 

DYNAMIC  PLATE  RESISTANCE.  When  the  steady  voltages  applied  to  a  triode  are 
such  as  to  cause  the  static  operating  point  to  be  within  a  curved  region  of  the  mutual 
Cor  other)  characteristic,  or  so  near  to  a  curved  portion  that  the  applied  alternating  voltage 
causes  the  triode  to  work  on  a  curved  portion  for  a  part  of  a  cycle,  more  accurate  re- 
sults will  be  obtained  from  the  series  expansion  by  evaluating  the  derivatives  at  the 
average  values  of  voltages  obtaining  during  the  cycle.  These  may  be  quite  different  from 

the  static  values.     The  plate  resistance  so  defined   ( —  =  —  )    is    called  the 

\rp       <ze&J  average/ 

dynamic  plate  resistance;  see  eq.  (12).  All  other  derivatives  should  likewise  be  evaluated 
at  the  dynamic  operating  point,  if  there  is  considerable  change  in  the  direct,  or  average, 
current  when  the  signal  is  impressed. 

The  dynamic  parameters  are  quite  tedious  to  compute  but  their  values  may  be  approx- 
imately measured  on  an  impedance  bridge  if  the  amplitude  of  the  applied  voltage  is  the 
same  as  that  of  the  signal  voltage.  The  resistance  measured  on  the  bridge  is  called  the 
effective  plate  resistance  and  for  small  or  medium  voltages  is  equal  to  the  dynamic  plate 
resistance.  When  they  are  not  equal  the  dynamic  resistance  is  the  value  to  be  used  in 
evaluating  the  series;  the  effective  value  must  be  used  if  it  is  desired  to  express  all  the 
fundamental  current  in  one  term. 

VARIATION  OF  AMPLIFICATION  FACTOR.  If  the  variation  of  /i  must  be  con- 
sidered (it  always  should  be  in  pentodes)  the  first  term  (ci),  Table  1,  is  unchanged  but  the 
second  term  (02)  is 


which  shows  the  increased  distortion  arising  from  the  variation  of  ju. 

22.  ANALYSIS  FOR  MULTI-ELECTRODE  TUBES 

The  above  method  of  analysis  can  be  extended  to  tetrodes,  pentodes,  etc.  If  voltage 
is  introduced  into  only  one  control  circuit,  and  if  the  other  grids  have  low  impedance,  the 
analysis  as  given  is  sufficient  for  any  case.  If,  however,  there  are  impedance  drops  in 
several  of  the  electrode  circuits  each  must  be  analyzed  in  the  same  manner.  The  equations 
are  very  complicated  and  not  of  sufficiently  general  interest  to  be  included  here  (see 
High-Frequency  Alternating  Currents,  McHwain  and  Brainerd,  John  Wiley  &  Sons,  Appen- 
dix A,  for  complete  development). 

23.  METHOD  OF  SUCCESSIVE  APPROXIMATIONS  * 

The  calculation  of  the  coefficients  in  the  power  series  expansion  of  a  vacuum  tube  and 
its  associated  circuit  becomes  extremely  complicated  for  terms  higher  than  the  third  term 
and  for  analyses  of  multi-electrode  tubes.  Thus  the  analysis  of  circuits  containing  vacuum 
tubes  by  means  of  the  Taylor's  series  expansion  of  the  tube  characteristics,  although 
theoretically  applicable  to  all  circuits,  is  applicable  in  practice  only  to  those  circuits  for 
which  the  series  converges  very  rapidly.  The  factors  affecting  the  convergence  are  (1)  the 
amplitude  of  the  applied  voltages  and  (2)  the  sharpness  of  the  curvature  of  the  character- 
istic in  the  operating  range,  that  is,  the  magnitude  of  the  higher  derivatives  of  the  char- 
acteristic. Thus  for  high  level  voltage  amplifiers,  power  amplifiers,  large  signal  detectors 
(including  linear  detectors),  and  most  oscillators,  the  treatment  by  Taylor's  series  is 
practically  useless  except  possibly  for  qualitative  analysis. 

In  such  cases  the  method  of  successive  approximations  is  sometimes  useful.  Equations 
of  the  form  ,^  , 

2)  (23a) 

s)  (236) 

are  written  for  the  total  number  of  currents  and  voltages  involved.f 

*  Articles  23  and  24  were  contributed  by  Dr.  Carl  C.  Chambers. 

f  This  method  is  applicable  to  certain  types  of  discontinuous  functions  such  as  those  found  in  dealing 
•with  gas-filled  tubes. 


5-46 


ELECTRIC   CIRCUITS,  LINES,   AND  FIELDS 


For  the  triode  with  voltages  in  both  the  plate  and  grid  the  equations  become 

ip  =  Fi({e'g  -  ig[  X  z,]},     [e'9  -  ip[  X  zj})  (24o) 

ig  «  ft({fl'*  -  «  X  zj},     {e'p  -  ip[  X  zp]})  (246) 

where  the  symbols  have  the  same  meaning  as  in  the  previous  section  on  the  Taylor's  series 
expansion  of  vacuum  tubes. 

The  values  of  the  resulting  currents  are  estimated  and  called  ipo  and  %)•  These  can  be 
obtained  from  oscillograms  of  a  similar  vacuum  tube,  from  the  approximate  solution  by 
means  of  the  Taylor's  series  expansion,  or  simply  from  an  intuitive  guess.  Although  the 
accuracy  of  the  estimate  of  the  value  of  ipo  and  igQ  is  not  theoretically  important,  the  labor 
involved  is  directly  dependent  upon  this  accuracy. 

Having  ipQ  and  igQ,  the  first  approximation  is  calculated  by  substitution  of  these  values 
in  the  right-hand  members  of  (24a)  and  (246),  giving 

*pi  =  Fi({St  ~  »«o[  X  zj}f     {e'p  -  ipot  X  zp]}) 
igi  =  Fi({efg  -  igo(  Xzg]},     {e'p  -  ipQ[  X  zp]}) 

The  second  and  higher  order  approximations  are  obtained  by  substituting  the  preceding 
approximation  in  the  right-hand  members  of  eqs.  (24). 

In  making  these  successive  substitutions  it  is  necessary  to  write  the  preceding  approxi- 
mation in  separate  sine-wave  terms  in  order  to  evaluate  igr[  X  zj  and  ipr[  X  zp].  This 
can  be  done  by  any  of  the  various  methods  for  harmonic  analysis  (see  Fisher-Hinnen, 
Method  and  Wave  Analysis,  article  1) .  When  zp  and  zg  are  selective  impedances,  the  only 
harmonic  components  of  the  current  of  importance  for  purposes  of  substitution  are  those 
for  which  zp  and  ze  are  not  essentially  zero.  For  this  reason  this  method  of  analysis  is 
especially  applicable  to  class  B  and  class  C  r-f  amplifiers,  oscillators,  and  frequency 
multipliers,  where  the  plate  impedances  are  highly  selective. 

The  functions  FI  and  F%  can  be  in  any  form  in  which  they  are  completely  specified  over 
the  entire  operating  range  of  the  applied  voltages.  Thus  any  analytical  expression  for 
the  tube  currents  or  any  complete  set  of  curves  is  sufficient  for  use  in  the  above  method  of 
analysis. 


24.  HARMONIC  ANALYSIS  OF  THE  CURRENT  FOR  A  SINUSOIDAL 

APPLIED  VOLTAGE 

In  many  cases  the  performance  of  a  non-linear  circuit  can  be  predicted  from  a  knowledge 
of  the  harmonic  content  of  the  current  when  a  sinusoidal  voltage  is  impressed  in  the  cir- 
cuit. The  usual  discussion  of  the 
merit  of  a  non-selective  amplifier  as- 
sumes a  sine-wave  excitation,  to  avoid 
the  complexity  introduced  by  the 
presence  of  the  many  beat  terms 
otherwise  present. 

When  the  relation  between  the  in- 
stantaneous impressed  voltage  and  the 
instantaneous  current  is  given  it  is  of 
course  possible  to  plot  the  wave  form 
of  the  current.  This  may  then  be 
analyzed  by  any  of  the  usual  methods 
of  wave  analysis. 

When  the  input  voltage  varies  sinu- 
soidally,  the  output  current  will  be 
periodic,  having  a  fundamental  peri- 
odicity equal  to  that  of  the  input  volt- 
age. In  resistive  circuits  the  relation 
between  the  input  voltage  and  the 
output  current  is  independent  of  time, 
so  that  the  output  current  will  pass 
from  maximum  to  minimum  and  from 
minimum  to  maximum  over  the  same 
path,  in  fact,  over  the  characteristic 
curve.  (The  term  characteristic  is 


FIG.  5.    Tube  Characteristic  and  Sine- wave  Input 


applied  in  its  original  sense  to  the  instantaneous  relation  between  input  voltage  and  output 
current).     It  follows  that,  if  a  complete  period  of  the  plate  current  is  T  seconds, 


HARMONIC  ANALYSIS  OF  THE  CURRENT      5-47 

=  i(T  —  t)  when  the  origin  of  time  is  chosen  at  the  instant  when  the  current  is  a  max- 
imum. This  form  of  symmetry  insures  that  when  the  output  current  due  to  a  sinusoidal 
input  voltage  is  written  in  the  form 

i  =  Jo  +  Ii  cos  (««  +  ]8i)  -h  Ii  cos  (2otf  -f  &)  +  (25) 

the  phase  angles  /?i,  £2,  etc.,  will  be  zero. 

Several  methods  of  analysis  are  used  to  obtain  the  values  Jo,  Ii,  etc.  The  one  of  broadest 
application  is  the  variation  of  the  Fisher-Hinnen  method  applicable  to  such  a  characteristic 
curve  instead  of  to  the  wave  itself.  The  analysis  by  this  method  can  be  stated  as  follows : 
consider  the  zero  point  of  the  sinusoidal  input  voltage  variation  to  be  the  zero  of  the  base 
line  of  the  characteristic,  and  consider  the  scale  along  this  base  line  to  be  such  that  the 
peak  value  of  the  input  voltage  is  unity.  Then  i(e)  is  the  relation  given  by  the  mutual  char- 
acteristic where  e  is  the  input  voltage  which  on  this  scale  varies  from  —  1  to  +1  and  i  is 
the  corresponding  output  current  (see  Fig.  5).  Then  the  /'s  to  the  second  approximation 
in  the  Fisher-Hi r>nen  method  of  analysis  are  given  by 

11  =  1/2[*U)  -*(-!)] 

12  =  1/4  [i(l)  -  2t(0)  +**(-!)] 

h  =  1/6  [*(D  -  2*(0.5)  +  2i(-0.5)  -  *(-D] 
I4  =  i/8[»(l)  -  2*(0.707)  +  2i(0)  -  2i(- 0.707) 


In  =    1   \iW-X  (cos 
2n  L  \w 


The  J's  to  the  fourth  approximation  in  the  Fisher-Hinnen  method  are  then  given  by 

I'n   =    In  -   I2n  -   Ifa 

The  d-c  component  Io  is  given  to  the  second  approximation  by 

Jo  _  Vl  [i(0)  +  ^+;("1}  +  i(0-707)+;("°-707)] 

Another  method  (see  D.  C.  Espley,  Proc.  I.R.E.,  Vol.  21,  1439  [1933])  has  the  advantage 
that  the  points  along  the  input  voltage  axis  of  the  mutual  characteristic,  at  which  the 
current  is  evaluated,  are  equally  spaced.  This  analysis  using  three  points  gives  results 
identically  the  same  as  the  values  for  "  Ii  and  /2  given  by  the  Fisher-Hinnen  method. 
When  five  points  are  used  the  I's  become 

Jo  =  1/6  fr'U)  +  2t(0.5)  +  2t(-0.5)  +  *'(-l)] 

Ii  =  1/«  [*(D  +  *(0.5)  -  i(-0.5)  -  t(-l)] 

I2=  1/4  [t(l)  -  2i(0)  +*(-!)] 

Is  =  Ve  WD  -  2t(0.5)  +  2i(-0.5)  -  *(-!)] 

/4  =  1/12  [*(!)  -  4i(0.5)  +  6i(0)  -  4£(-0.5)  +  *(-!)] 

For  the  values  of  the  I's  using  seven  equally  spaced  points  see  the  original  paper  referred 
to  above. 

In  the  special  case  when  the  even  harmonics  are  balanced  out  as  in  "back  to  back" 
amplifiers  (pushpull  class  A,  class  B  audio,  and  class  AB  amplifiers,  see  Amplifiers,  Section 
7)  the  mutual  characteristic  becomes  like  that  shown  in  Fig.  6.  I0  is  balanced  out  of  the 
output  current  as  well  as  the  even  harmonics  so  that  i(e)  =  —i(—e}.  For  such  amplifiers 
the  analysis  by  the  Fisher-Finn  en  method  gives  for  the  remaining  I's 

Ii  =  *(D 

/s  =  1/3  Pd)  -  2^(0.5)] 

Is  =  l/5  [i(l)  -  2i(0.809)  +  2t(0.309)] 

I7  -  1/7  [t(l)  -  2i(0.901)  +  2i(0.624)  -  2»(0.222)I 


As  before,  a  better  approximation  for  In  is  obtained  by  subtracting  I&n  from  the  cal- 
culated In. 


5-48 


ELECTRIC  CIRCUITS,  LINES,   AND  FIELDS 


When  this  symmetry  exists  the  method  of  Espley  described  above  reduces  to  the  Fisher- 
Hinnen  method  when  the  measurements  are  made  at  only  five  points  (for  this  special 
case  only  two  points  need  actually  be  measured),  and  the  seven  measured  point  analysis 
of  Espley  reduces  to 

Ii  =  1/320  [167i(l)  4-  252i(0.667)  -  45^•(0.333)] 

h  =  Vi28  [45i(l)  -  36^(0.667)  -  63i(0.333)] 

IB  =  8i/640  [>•(!)  -  4^(0.667)  +  5z(0.333)] 

Here  because  of  the  symmetrical  character  of  the  curve  only  three  points  actually  need 
be  measured. 

Mouromtseff  and  Kozanowski  (Proc.  I.R.E.,  Vol.  22,  1090  [1934])  give  a  somewhat 
simpler  method  to  calculate  up  to  the  eleventh  harmonic  in  the  case  of  a  symmetrical  curve 
such  as  Pig.  6.  The  straight  line  aob  is  drawn  intersecting  the  curve  at  the  point  of  maxi- 
mum input  voltage,  that  is,  e  —  1  in  the  notation  used  above.  Then,  instead  of  measuring 
i(e),  Ai(e)  is  measured  where  Ai(e)  is  the  difference  in  current  between  the  curve  and  the 
line  006  and  is  taken  positive  when  the  line  006  is  above  the  curve.  This  gives  for  the  Pa 
in  the  order  in  which  they  are  to  be  calculated 

IB  =  0.4^(0.309)  -  At(0.809)] 

J3  =  0.4475[Ai(0.309)  +  Ai(0.809)]  +  0.333Ai(0.5)  -  0.578Ai(0.866)  -  0.5I6 

IT  =  0.4475[Ai(0.309)  +  At(0.809)]  -  Ji  +  0.5J5 

J9  *  /3  -  0.667A*(0.5) 

In  =  0.707 A£(0.707)  -  J3  -f  Is 

Ji  =  i(l)  -  Ij  +  IB  -  IT  +  Is  -  In 

Any  of  the  above  methods  can  be  used  for  the  calculation  of  the  I's  in  eq.  (25)  when 
the  input  voltage  is  sinusoidal.  The  per  cent  amplitude  for  any  given  harmonic  is  then 
In/Ii  for  an  input  voltage,  the  peak  value  of  which  is  taken  e  =  1.  The  complete  equation 
of  the  curve  from  e  —  —I  to  e  —  +1  can  be  written  in  the  form 

i(e)  =  IQ  +  lie  +  h  cos  2(cos~1e)  +  Is  cos  3(cos~1e)  +  •  •  • 

This  expression  for  the  output  current  is  unique;  that  is,  for  each  value  of  e  between 
—  1  and  + 1  this  equation  gives  the  corresponding  instantaneous  plate  current  provided 

that  the  conditions  prescribed  at  the 
beginning  of  this  section  are  fulfilled, 
chiefly,  that  the  load  is  essentially  a 
constant  resistance  over  the  operating 
frequencies  and  voltages.  This  expres- 
sion can  be  used  in  several  ways  although 
calculations  of  the  operation  for  input 
voltages  other  than  a  simple  sinusoid  of 
amplitude  unity  on  the  scale  of  e  are 
extremely  complex  analytically. 

The  calculation  of  the  distortion  and 
output  by  any  of  the  above  methods  for 
a  sinusoidal  input  voltage  having  an 
amplitude  of  a  fixed  fraction  of  e  can  be 
made  by  means  of  this  expression  for 


Output 


-1 


Current 


&      — *    Input  Voltage 


i(e)  without  again  using  the  curve. 
Formulas  can  be  developed  for  this  pur- 
pose using  any  of  the  above  methods  of 
analysis  as  a  basis  giving  the  I's  corre- 
sponding to  an  input  voltage  of  frac- 
'Back  tional  amplitude. 

This  method  of  analysis  can  generally 
be    used    when    the    resistive    load    is- 
coupled  into  the  non-linear  circuit  by  a  transformer.    For  if  the  transformer  is  "perfect" 


FIG.  6.     Mutual  Characteristics  of  Two  Tubes 
to  Back" 


the  relation  between  voltage  and  current  is  the  same  as  for  a  resistance  load  (see  Trans- 
formers, Section]  6) ;  in  other  words  the  transformer  offers  a  resistive  impedance  to  the 
circuit.  Practical  transformers  are  nearly  perfect  enough  to  assume  that  their  only  effect, 
is  to  multiply  (or  divide)  the  load  resistance  by  the  square  of  the  turn  ratio. 


ELECTROMAGNETIC   RADIATION  5-49 


25.  INPUT  IMPEDANCE  OF  A  TRIODE 

Neglecting  the  leakage  resistances  between  electrodes  the  equivalent  circuit  of  a  triode 
with  grid  negative  is  shown  in  Fig.  7.    The  equivalent  input  resistance  is 

C3- 

(26o) 


the  equivalent  input  reactance  by 

f  wVCCfl  +  C'sXr2  +  a?) A  +  B(r*  +  re2  +  rr. 


)  \ 
C3)  (rp2  +  rrp)  -  wrp2^(2A  +  C32)     f 


Tpl 

Cz)}}\ 


where  A  «  dC2  +  dC3  +  C2C3,  and  B  =  Ci  -f  C2  -f  ^C3. 

Both  the  numerator  and  denominator  of  each  of  these  expressions  contain  negative 
terms ;  thus  either  can  be  positive  or  negative  depending  on  the  circuit  parameters.  If  the 
value  of  the  reactance  becomes  positive  it  indicates  that 

the  equivalent  input  reactance  is  inductive.    If  the  value  1  G?d       1L      iplatei 

of  the  expression  for  resistance  becomes  negative,  it  in- 
dicates that  the  real  part  of  the  input  impedance  of  the 
tube  is  an  equivalent  negative  resistance  and  is  supplying 

instead  of  absorbing  power.    The  input  resistance  will         t  

become  negative  only  for  certain  values  of  inductive  load;  '         Filament 

cf.  eq.  (26a).     When  this  condition  exists,  the  tube  will    -n,      -     0.     .._   ,  „     .     .     .  ~. 

,  i  .         i    .  ,   .  r        ..  TJ  f        FIG.  7.    Simplified  Equivalent  Cir- 

always  supply  a  greater  plate  current  than  it  would  for  cuft  Of  a  Tnode 

the  same  grid  voltage  and  a  non-reactive  load.    When 

this  occurs  the  tube  is  said  to  be  regenerating  (see  Section  S) .  If  the  negative  input  resist- 
ance exactly  equals  the  external  resistance  the  total  resistance  of  the  circuit  is  zero  and 
current  can  flow  without  a  driving  voltage;  in  such  a  case  the  tube  is  said  to  oscillate  (see 
Section  7,  oscillators). 

When  the  grid  voltage  is  positive,  grid  current  flows  and  the  grid  resistance  is  defined 
by  lA*£  =  dis/deg.  This  is  practically  infinite  when  the  grid  is  negative  but  may  be  quite 
small  when  the  grid  is  positive;  in  this  latter  case  the  interelectrode  impedances  are  usually 
negligible  in  comparison  to  it  so  that  the  input  impedance  of  the  triode  is  simply  re. 

BIBLIOGRAPHY 

Byerly,  W.  E.,  Elements  of  Integral  Calculus,  p.  332.    G.  E.  Stechert  and  Co. 

Carson,  J.  R.,  A  Theoretical  Study  of  the  Three-Element  Vacuum  Tube,  Proc.  I.R.E.,  April  1919. 

Steinmetz,  C.  P.,  Engineering  Mathematics,  Chapter  VI.    McGraw-Hill. 

Mcllwain,  K.,  and  J.  G.  Brainerd,  High-frequency  Alternating  Currents,  Chapter  VI.    John  Wiley. 

Terman,  F,  E.,  Radio  Engineering.    McGraw-Hill  (1932). 

Llewellyn,  F.  B.,  Operation  of  Thermionic  Vacuum  Tube  Circuits,  5.S.T.J".,  Vol.  V,  433  (1926). 

Barrow,  W.  L.,  Contribution  to  the  Theory  of  Non-linear  Circuits  with  Large  Applied  Voltages,  Proc. 

I.R.E.,  Vol.  22,  No.  8,  964  (August  1934). 
Byerly,  W.  E.,  Elementary  Treatise  on  Fourier  Series  and  Spherical,  Cylindrical,  and  Ellipsoidal  Har~ 

monies.    Ginn  (1902). 
Jahnke.  E,,  and  F.  Emde,  Tables  of  Functions  with  Formulae  and  Curves,  2nd  Ed.,  p.  242.   B.  G.  Teubner,, 

Berlin  (1933). 


ELECTROMAGNETIC  RADIATION 

By  Knox  Mcllwain 

All  forms  of  wireless  or  radio  transmission  depend  on  the  fact  that  electromagnetic 
energy  is  radiated  from  any  wire  in  which  a  varying  current  flows.  The  current  in  the- 
wire  sets  up  magnetic  and  electric  fields,  in  which  it  is  usually  assumed  that  the  energy 
associated  with  the  current  flow  is  stored;  a  portion  of  this  energy,  however,  is  not  stored, 
but  continually  travels  away  from  the  wire,  or  is  radiated.  Radiation  is  a  phenomenon. 
which  is  totally  negligible  at  low  frequencies. 


5-50  ELECTRIC   CIRCUITS,  LINES,   AND  FIELDS 

26.  MAXWELL'S  EQUATIONS 

There  are  two  well-known  experimental  relations  between  magnetic  and  electric  fields. 
The  two  fields  are  interdependent,  a  change  in  one  always  being  accompanied  by  a  change 
of  the  other.  The  two  laws  are: 

1.  Whenever  the  net  magnetic  flux  linking  any  closed  loop  changes,  an  electromotive 
force  is  set  up  in  the  loop.    If  4>  is  the  net  magnetic  flux  linking  the  loop,  then  (Faraday's 
law)  the  electromotive  force  is 

•--2 

2.  The  resultant  ma|netomotive  force  (in  ampere-turns)  acting  around  any  closed  loop 
is  equal  to  the  rate  of  flow  of  electricity  through  the  surface  bounded  by  the  loop  plus  the 
time  rate  of  change  of  electric  flux  through  this  surface.    If  i  =  dq/dt  is  the  rate  of  flow  of 
electricity  (total  conduction  current  in  amperes)  and  ^  the  electric  flux  through  the  surface 
then  the  magnetomotive  force  is 

dd 

m  =  ^  +  —  (2) 

at 

Both  these  relations  are  such  that  the  force  expressed  by  the  left  side  of  the  equation 
bears  a  right-hand-screw  relation  to  a  positive  increment  of  the  right-hand  side.  They 
apply  to  any  medium  whatever,  to  conductors,  dielectrics,  ferromagnetic  substances,  etc. 

By  applying  these  equations  to  an  elementary  volume  in  space,  Maxwell's  laws  of 
electromagnetism  are  developed.  Stated  in  vector  notation  these  are  (all  quantities  are 
expressed  in  mks  units) 

dH  An 

—  ju  — -  =  curl  £  (4) 

at 

From  these  the  continuity  equations  immediately  follow; 

div  u  —  0  (5) 

div  B  =  0  (6) 

where  u  is  the  total  current  density  in  amperes  per  square  meter  and  B  the  magnetic  flux 
•density  in  webers  per  square  meter. 

Stated  hi  the  ordinary  component  form  these  equations  are: 

(3a) 
(36) 
(3c) 
(4a) 
(46) 


dDx      ,  ,        y      ,  ,  n  ,K  . 

(5a) 

-where  <r  is  the  conduction  current  density  in  amperes  per  square  meter  and  D  the  electric 
.flux  density  in  coulombs  per  square  meter,  and 

^  +  ^2  +  ^  =  0  (to) 

dx         dy         dz 

'Only  fields  which  satisfy  these  relations  are  possible,  and  these  equations  may  be  used  to 
-determine  the  necessary  form  for  particular  fields. 


PROGRESSIVE  PLANE  WAVES  5-51 

The  energy  flow  associated  with  the  electromagnetic  wave  is  found  by  developing 
Poynting's  vector 

P  =  e#sin(£,  H)  (7) 

which  is  a  vector  perpendicular  to  8  and  H,  whose  magnitude  depends  on  the  product  of 
e  and  H  and  the  sine  of  the  angle  between  them  (measured  counterclockwise  from  £  to  H)  . 
The  integral  of  Poynting's  vector  over  a  surface  gives  the  rate  at  which  energy  flows 
through  the  surface. 

WAVE  EQUATION.  In  an  isotropic  insulating  medium  no  conduction  current  can 
flow,  since  7  (eq.  3)  is  zero.  Elimination  of  H  between  eqs.  (3  and  4),  and  similarly  for  £, 
gives  the  wave  equations 

*?_  J_  /*?  +  ££  +  d^ 
df  ~  V.K  W  +  w  +  a? 


Solution  of  these  equations  shows  that  each  component  of  electric  or  magnetic  field  must 
have  the  form 


where  v  =  ±l/Vju5.  These  functions  represent  incident  and  reflected  waves  (see  p.  5-23) 
traveling  at  the  velocity  v  (  =  3  X  10s  meters  per  sec  in  vacuum  and  close  to  that  in  air)  . 
The  wavelength  is  defined  as  the  distance  traveled  in.  one  period. 

27.  PROGRESSIVE  PLANE  WAVES 

The  simplest  form  of  wave  mathematically  is  one  which  travels  along  the  x  axis,  say, 
and  in  which  it  is  assumed  that  none  of  the  fields  vary  with  y  or  2,  In  such  case  (by 
Maxwell's  laws) 

e*  =  0  (lla) 

Sy   =    8y  COS       CO  (t  -   -  J  (116) 


e,  ==  e*  cos 
Hx  =  0  (lid) 


pr     A/_±.  $? 

jti  2    —      V  —  C-y 

in  which  co  =  2-n-/.    Then  «[*  -  (x/a)]  =  wt  -  2irx/\. 

In  the  general  form  the  loci  of  the  vectors  e  and  JT  are  ellipses,  so  that  the  plane  wave 
above  is  said  to  be  elliptically  polarized.  If  ey  =  Bz  and  dz  —  ir/2  or  37T/2  the  ellipse  reduces 
to  a  circle  and  the  wave  is  said  to  be  circularly  polarized. 

If  either  sv  or  ez  is  zero,  or  if  dz  —  0  or  TT,  the  ellipse  reduces  to  a  straight  line,  in  which 
case  the  wave  is  called  a  plane-polarized  wave.  The  electric  energy  per  meter3  of  such  a 
wave  at  any  point  in  the  field  is  2?e2/2,  and  the  magnetic  energy  per  meter3  is  jufir2/2. 
It  follows  that  for  a  progressive  plane-polarized  wave  at  every  point  the  total  energy  is 
always  half  electric  and  half  magnetic,  but  its  value  varies,  of  course,  with  position  and 
time.  The  total  energy  (W)  of  the  field  is 


W  =  We  +  Wm  =  K&dr  (12) 


where  dr  is  an  element  of  volume.    The  average  energy  is  Wi  —  Xe2/2T  and  the  average 
rate  at  which  energy  passes  through  a  meter2  in  the  YZ  plane  is 


5-52  ELECTRIC  CIRCUITS,  LINES,  AND  FIELDS 


28.  FIELDS  DUE  TO  A  CURRENT  IN  A  WIRE 

If  a  current  i  flows  in  a  short  length  of  wire  (see  Fig.  1)  located  in  free  space  and  assumed 
at  the  origin  of  coordinates  and  coincident  with  the  z  axis,  the  fields  at  any  point  P(x,  y,  z) 
are 

-  -  *& 

*- 

(WC) 


J-T- 

dydt 


Hz  =  0  (14/) 

where  w  ^  -fi  [  t  —  «-  )  ;  in  this  expression  d  is  the  distance  from  the  origin  to  the  point, 

d     \        v 

and  Fi(t)  expresses  the  distribution  of  charge  in  the  wire,  so  that  F(t)  =  idl,  where  81  is 
the  length  of  the  wire  (assuming  uniform  current  for  the  differential  length  81)  . 

If  the  current  is  assumed  sinusoidal,  so  that  i  =  V2I  sin  (ut  +  90°)  amperes,  at  any 
point  distant  from  the  wire  (so  that  d  ^>  X/2?r)  ,  the  fields  are 

cos  6  cos  (ut  —  pd)  volts  per  meter  (15a) 


,^  „ 
150  aA 


COs  6  cos  (w£  —  (3d)  ampere-turns  per  meter 


where  6  is  the  angle  of  elevation  of  the  point  P  from  a  plane  perpendicular  to  the  wire. 
All  lengths  are  expressed  in  meters. 

The  direction,  of  the  electric  field  is  perpendicular  to  d  and  in  the  plane  formed  by  d 
and  the  axis  of  the  wire.    The  magnetic  field  is  perpendicular  to  d  and  the  electric  field. 
Such  an  elementary  wire  is  called  an  electric  doublet.    The  fields  due  to  a  long  length 
of  wire  I  can  be  obtained  by  integration  of  the  elementary  length  (see  Antennas,  Section  6.) 
The  average  rate  of  flow  of  energy  per  square  meter  through 
an  area  perpendicular  to  d  is  (in  watts) 

and  the  total  power  radiated  (in  watts)  is 


FIG.  1.     Elementary  Radi-  p_  _       ?!- 

ator  rr  ~        \2 

The  radiated  power  for  a  given  current  varies  directly  as  1/X2  or  as  the  square  of  the 
frequency.  For  a  given  total  charge,  or  a  given  voltage  difference  between  the  ends  of  the 
antenna,  the  radiated  power  varies  as  1/X*  or  as  the  fourth  power  of  the  frequency. 

29.  REFLECTION  AND  REFRACTION 

Whenever  an  electromagnetic  wave  meets  a  boundary  between  two  media  of  different 
dielectric  or  magnetic  properties,  there  is  a  change  in  the  fields  and  frequently  the  wave 
splits  into  two  waves,  one  of  which  is  reflected  back  into  the  first  medium,  the  other  is 
refracted  into  the  second  medium. 

CONDITIONS  AT  THE  BOUNDARY.  The  following  proposition  is  assumed:  In  two 
media,  the  components  of  the  electric  (or  magnetic)  intensity  tangent  to  the  surface  sepa- 
rating those  media  are  equal  (in  magnitude  and  direction)  at  this  surface.  It  follows  from 


REFLECTION  AND  REFRACTION 


5-53 


this  that  the  normal  component 
of  the  surface,  that  isr  UN  —  u^ 

Similarly 


of  the  total  current  density  is  the  same  on  both  sides 
or  for  insulating  media 

DN  —  D^N 


The  general  expression  for  a  plane  wave  as  far  as  £  is  concerned  is: 


£x  COS  OJ  I  t 


8y  J 

co  f  i  -  -j  -f  5* 


(18) 
(19) 

(20<z) 
(206) 
(20c) 


where  5y  and  5Z  are  constants  and  &x  is  the  maximum  value  of  the  X  component  of  £. 

TWO  ISOTROPIC  DIELECTRICS.  Consider  two  media  separated  by  a  plane  sur- 
face, and  assume  that  a  plane  wave  in  medium  1  impinges  on  this  surface  (see  Fig.  2). 
Assume  that  the  total  electric  intensity  £  in  the  first  medium  is  composed  of  two  parts 
£1  and  £'i;  then 

£  =  £1  +  £'i  (21) 

£1  will  be  called  the  incident  wave  (see  also  p.  5-23)  and  £'i  the  reflected  wave;  £'i  will  have 
components  similar  to  those  of  eq.  (20)  denoted  by  prunes.  Note  that,  3/1  being  undeter- 
mined, what  is  called  the  reflected  wave  is  not  restricted  to  travel  in  a  direction  directly 
opposite  the  incident  wave,  as  is 
the  case  on  transmission  lines. 

Assume  that  the  resultant  elec- 
tric intensity  in  the  second  medium 
is  £2,  called  the  refracted  wave, 
with  similar  components  identified 
by  the  subscript  2. 

Applying  Maxwell's  equations 
at  the  boundary,  &I-QI  —  /3'ifl'i  = 
182^2  or  coi  =  oo 'i  =  co2»  so  that  all 
waves  have  the  same  frequency. 

Also  the  incident,  reflected,  and 

refracted  waves  all  have  wave  nor-     '  ~Y  Origin  is  at  0 

mals  (perpendiculars  to  the  wave  FIG.  2.    Wave  Normals 

front)  in  the  same  plane.    Similarly 

if  0  is  the  angle  between  the  wave  normal  and  the  normal  to  the  boundary  then  « 
or  the  angle  of  incidence  is  equal  to  the  angle  of  reflection.    Also 


Normal  to 

Interface  Between 

Two  Media 


Normal  to  Wave  Front 
of  Incident  Wave,  or 
Incident  Wave  Mojcmal 


sin  (£2 


:  -  =  f\ 


(22) 


where  TJ  is  the  relative  index  of  refraction  of  medium  2  with  respect  to  medium  1. 

The  magnetic  intensities  of  the  various  waves  are,  in  magnitude  (from  Maxwell's 
equations),  HI  =  V ' K\l Vi£i,  Hfi  ~  v'JSi/juie'i,  and  #2  —  V Kz/^fy.  All  these  are  vectors 
in  the  wave  fronts  such  that  £1,  Hi,  and  pi  are  mutually  perpendicular  in  such  orientation 
that  rotation  from  £1  to  HI  (through  the  smaller  angle)  would  move  a  right-handed  screw 
in  the  direction  of  p. 

Normal  Components.  It  is  convenient  to  split  the  electric  and  magnetic  intensities  into 
two  components,  one  normal  to  the  plane  of  incidence  (not  the  interface  between  the  media) 
and  one  in  the  plane  of  incidence.  Maxwell's  equations  apply  to  each  separately,  since 
the  equations  are  assumed  linear  (there  has  been  some  indication,  the  intermodulation  of 
waves  in  space,  that  the  equations  are  not  entirely  linear) .  The  normal  components,  at 
the  interface  in  terms  of  the  incident  wave,  are 


Z'IN  ,~,  cos  fa  VT^T  —  cos  c})2 ' 

/8'iN  =  


822V 
— 


2  cos  <£i ' 


+  cos 


(23a) 


(236) 


5-54 


ELECTRIC   CIRCUITS,  LINES,   AND  FIELDS 


Since  cos  fa  =  Vl  —  sin.2  <fe  and  since  sin  0i/sin  fa  =  T\  (the  index  of  refraction)  ,  then 
cos  02  =  Vl  —  sin2  0i/V  Hence,  if  sin  0i  >  77,  cos  $2  will  be  imaginary.  This  is  the  case 
of  total  reflection  (see  below)  .  The  angle  of  incidence  for  which  sin  0!  =  77  is  called  the 
critical  angle, 

Parallel  Components.  The  parallel  components  at  the  interface,  in  terms  of  the  inci- 
dent wave,  are 

S'IP       e'iP  /B/          cos  0i 
-  —  =  =  —  /5'ip  = 

£lP  8lP 


82P          82P    /s 

-  =  =—  /Q2P 

8lP          81P 


COS  01  VfJLlKz  +   COS  02 
2  COS  01  VfJL^Ki 

- 


(246) 


cos  0].  V  Ml#2  +  COS 

Case  of  Total  Reflection.    If  cos  02  is  imaginary  let  cos  02  =  jn^  where  n^  is  the  absolute 
value  of  the  imaginary.    Substituting  this  in  eqs.  (23)  and  (24)  ,  the  normal  components  are 

cos  0i 


COS  01 


(25) 


and  the  parallel  components  are 


cos  0i 


Refracted 
Beam 


X  There  is  a  disturbance  in  the  second  medium  but  no 
energy  is  transmitted,  so  that  the  wave  is  said  to  be 
totally  reflected. 

Energy  Relations.     To  compare  the   energy   in  the 
Incident    incident,  reflected,  and  refracted  waves  determine  the 
Beam     rate  at  which  energy  is  delivered  by  the  incident  wave 
to  a  certain  area  of  the  surface  separating  the  two  media, 
and  the  rates  at  which  it  leaves  this  area  in  the  reflected 
and  refracted  waves  (see  Fig.  3) . 


The  respective  energies  are 

Incident  beam:     cos  0i  dS  V—  8i2  =  \/—  8i2d£i 
^  u  v 


Reflected  beam:     cos  0i  dS 


f^"  S" 

A/—  e'i2  =   \~ 

V  * 


Refracted  beam:     cos  02  d>S 


—  822 


—       ~  822d>Si 


(27a) 


(276) 


(27c) 


i  cos  0i 
The  ratio  of  reflected  to  incident  energy  is  eV/ei2,  and  of  refracted  to  the  incident  energy 

- 


energy  is 


"^  —  (  =•  I  » 
KIM  Vei/ 


where  the  8's  are  evaluated  at  the  interface. 


cos  0i 

Rotation  of  Plane  of  Polarization.  If  the  incident  wave  is  a  plane-polarized  wave,  and 
the  plane  of  polarization  (which  contains  HI  and  hence  is  perpendicular  to  Ci)  makes  an 
angle  0i  with  the  plane  of  incidence,  then  the  angle  between  the  plane  of  polarization  and 
the  plane  of  incidence  of  the  reflected  wave  (0'i)  is 


and  for  the  reflected  wave 


tanfc 


(28) 


(29) 


1   -   02) 

When  0  <  &i  <  Tr/2,  the  plane  of  polarization  of  the  reflected  wave  is  rotated  toward  the 
plane  of  incidence,  while  that  of  the  refracted  wave  is  rotated  away  from  the  plane  of 
incidence. 


REFLECTION  AND  REFRACTION          5-55 

CASE  OF  A  CONDUCTING  MEDIUM.     If  a  medium  is  conducting,  its  conductivity 
is  not  zero,  and  by  Maxwell's  first  equation  (p.  5-50). 


T£  +  -j-  =  curl  H  (30) 

where  all  quantities  are  in  mks  units.    Assuming  £  to  be  a  harmonic  function,  it  can  be 
written  £  =  /  (xyz)  £/w*  and  d£/dt  =  /co£.    Hence  eq.  (30)  becomes 

-^  =  curl  H  (31) 

This  equation  is  similar  to  that  for  a  dielectric,  the  only  difference  mathematically  being 
that  the  coefficient  of  dt/dt  is  the  constant  K,  whereas  in  eq.  (31)  it  is  the  constant 
K  —  JT/W.  Hence  all  results  obtained  for  insulating  media  are  applicable  to  a  conducting 
medium  provided  the  K  used  in  the  former  case  is  replaced  by  K  —  jj/u. 

If  8  =  ~£ie~~ax*?(>Cj*t-px')i  -which  is  a  damped  plane  wave  propagated  in  the  X  direction 
with  velocity  co//3  and  attenuation  a.  per  unit  length,  then  for  this  to  be  a  solution  of 


./ry\  52e 

i  -  3  —  )  8  =  — ^ 

co  /  d^ 


(32) 


(wave  equation,  all  parameters  are  those  of  the  conducting  medium)  it  is  necessary  that 
a  =  2/ryfl.  For  good  conductors  y  is  large,  hence  a.  is  large  and  the  wave  is  highly  atten- 
uated —  good  conductors  are  poor  transmitters  of  electromagnetic  waves.  Indeed,  a 
perfect  conductor  (j  =  oo)  will  not  transmit.  These  results  hold  for  wavelengths  not 
near  the  visible-light  range  when  the  conductor  (called  in  this  case  a  reflector)  is  an  ordi- 
nary metal. 

REFLECTION  FROM  A  CONDUCTOR.  If  a  wave  traveling  through  an  insulating 
medium  (1)  strikes  the  (plane)  interface  of  an  adjacent  uniform  conducting  medium  (2) 
all  the  results  previously  obtained  for  the  case  of  two  insulating  media  are  applicable 
provided  Ki  is  replaced  by  K'z  —  jy/u,  where  K'%  now  represents  the  dielectric  constant 
of  the  conducting  medium.  Assuming  directions  shown  in  Fig.  2,  a  plane  wave  in  the 
insulating  medium  produces  in  the  conducting  medium  a  wave 

g2  =  e2eaVffa)*  ~  0'2  fr  sin  *'2  ~  2  cos  *'a)  +  *2  J  (33) 

where  a  is  a  positive  real  quantity  (z  is  negative  so  the  wave  is  damped)  and  to,  £'2,  ^'2, 
and  52  are  all  real.  The  quantity  w  is  2-rrf  where  /  is  the  frequency  of  the  incident  wave, 
and  7/2,  the  velocity  of  the  wave  in  the  conducting  medium,  is  w/P'z.  Furthermore, 
£'2  =  27r/X/2,  where  X'g  is  the  wavelength  of  the  wave,  and 


/   -          -  =  /04.x 

17   "  V2       fa        sin  0'2  ^     ; 

is  the  refractive  index  of  medium  2  with  respect  to  medium  1.  The  various  quantities 
a,  j372,  sin  ^'2,  cos  ^'2,  2/2,  and  if  are  all  functions  of  the  angle  of  incidence  <fo.  It  is  customary, 
when  not  otherwise  specified,  to  assume  that  quoted  values  of  a,  £'2,  3/2,  and  17'  are  for 
normal  incidence  (<fo  =  0).  To  compute  these  quantities  for  any  angle  of  incidence, 
the  following  procedure  may  be  used  (K  is  written  for  Kfz  and  subscript  N  refers  to  normal 
incidence;  T  to  27r/cj).  Calculate  or  measure 


Then  77'  can  be  found  from 


and  1/2,  P'z,  and  $'2  follow  from  eq.  (34).    The  following  are  of  importance: 

^/rK7z  only  when  jT  is  small  and  K\  =  1 
i  when  yT  ^>  1 


when  yT  ^>  1 
when  yT  ^>  1 


5-56 


ELECTRIC   CIRCUITS,  LINES,   AND   FIELDS 


The  ratio  of  energy  reflected  from  to  energy  incident  upon  the  interface  per  unit  time  is 
e  -  (2/V-yT  cos  <fr)  for  yT  large. 

As  7  approaches  oo,  this  ratio  approaches  unity,  showing  that  good  conductors  are  good 
reflectors.    For  normal  incidence  (<£i  =  0)  it  becomes 


and  in  general,  assuming 


,  it  is 
(m  -  ^ 


(ni  + 


+ 


It  may  be  noted  that  in  a  conducting  medium  neither  the  electric  nor  the  magnetic 
vector  lies  in  the  wave  front. 

The  preceding  discussion  has  assumed  K'%  a  constant,  whereas  it  is  sometimes  necessary 
(e.g.,  in  Heaviside  layer  studies)  to  consider  K'%  and  other  quantities  as  varying. 

BIBLIOGRAPHY 

Ramo,  S.f  and  J.  R.  WMnnery,  Fields  and  Waves  in  Modern  Radio.    John  Wiley  (1944). 
Skilling,  H.  H.,  Fundamentals  of  Electric  Waves.    John  Wiley  (1942). 
Stratton,  J.  A.,  Electromagnetic  Theory.    McGraw-Hill  (1941). 
Schelkunoff,  S.  A.,  Electromagnetic  Waves.    D.  Van  Nostrand  (1943). 
Slater,  J.  C.,  Microwave  Transmission.    McGraw-Hill  (1942). 


ELECTROMECHANICAL  SYSTEMS 

By  Knoz  Mcllwain 


Support 


Vibrations  in  one  dimension  occur  frequently  in  systems  made  up  of  solid  elements,  such 
as  a  spring  suspension,  a  pendulum,  or  a  rocking  lever.  They  occur  occasionally  in  fluids, 
an  example  being  the  transmission  of  sound  through  a  long  narrow  tube.  Alternating 
electric  currents  in  short  wires  will  be  shown  to  be  similar  analytically  to  unidimensional 
mechanical  vibrations.  Also  devices  are  in  common  usage  (microphones,  speakers,  etc.) 
which  convert  any  of  these  forms  of  vibration  into  any  other. 

Because  of  the  analytical  similarity  of  the  three  forms  of  vibration,  mathematical 
results  obtained  in  one  field  may  be  used  in  solving  problems  in  the  other  fields.  The 
mathematics  involved,  such  as  the  complex  notation,  has  been  applied  more  generally  to 
the  electrical  problem  than  to  the  others,  but  once  the  analytical  similarity  is  established 
advances  in  one  field  are  immediately  applicable  to  the  others. 

As  a  first  approximation  all  the  parameters  are  considered  constants.  This  is  reasonably 
true  in  many  electrical  circuits  but  is  not  generally  so  in  mechanical  systems.  However, 

the  variations  for  small  displacements  and  ve- 
locities such  as  occur  in  mechanical  systems  used 
in  acoustics  are  usually  negligible.  The  methods 
of  attack  available  when  the  variation  of  param- 
eters must  be  considered  are  given  in  articles 
16-24. 

DEFINITIONS.  The  number  of  independent 
variables  required  completely  to  specify  the 
motion  of  every  part  of  a  vibrating  system  is  a 
measure  of  the  number  of  degrees  of  freedom  of 
the  system.  When  only  one  variable  is  needed 
the  system  is  said  to  have  one  degree  of  free- 
dom; examples  of  such  systems  are  (Fig.  1):  a 
piston  moving  in  a  cylinder,  a  weight  hanging 
from  a  spring,  cylinder  rolling  on  a  plane  surface 
(no  slipping).  Systems  of  two  or  more  degrees 
of  freedom  are  exemplified  by  an  automobile  moving  on  a  plane  (two  degrees  if  no  skid- 
ding, three  if  skidding  occurs),  a  balloon  (three  degrees  if  there  is  no  spinning  of  the 
balloon),  an  airplane  (six  degrees  if  rotation  is  considered),  a  set  of  weights  connected 
by  springs  in  series  (Fig.  2)  which  has  as  many  degrees  of  freedom  as  there  are  springs; 
continuous  vibrating  systems  (waves  traveling  along  a  long  spring  or  stretched  string) 
have  an  infinite  number  of  degrees  of  freedom. 


Equilibrium 

•  Position  of 

M 

(a)  Simple  Mechanical  ( 

System 


FIG.  1. 


"&)  Electrical  System 
"Equivalent"  to 
Mechanical  System 
of  (a)    - 

Electrical  and  Mechanical  Systems 
of  One  Degree  of  Freedom 


ENERGY  OF  MECHANICAL  AND  ELECTRICAL  SYSTEMS      5-57 

Consider  a  system  of  one  degree  of  freedom  (Fig.  1)  ,  and  let  the  displacement  coordinate 
be  chosen  to  measure  the  distance  from  the  center  of  the  mass  M  to  the  equilibrium  posi- 
tion. Consider  that  the  spring  has  mass.  The  velocity  of  any  given  elementary  mass  of 
the  system  will  then  be  proportional  to  ds/dt(  =  s),  so  that  the  kinetic  energy  of  this 
elementary  mass  dm  will  be  */2  ki&dmi,  where  ki  is  a  constant  dependent  on  the  position 
of  the  mass.  Likewise  the  kinetic  energy  of  any  other  elementary  mass  dm*  will  be 
z,  etc.,  and  the  total  kinetic  energy  T  of  the  system  will  be  the  sum  of  these,  or 


T  -  i/zfh's2  dim  =  1/2  a*  fki  dmi  (1) 

where  the  integration  is  to  extend  over  the  entire  mass  composing  the  system.    But 

Jki  dmi  -  mm  (2) 

where  mm  is  a  constant,  called  the  generalized  mass.    Note  that  the  value  of  mm  depends 
on  the  k's,  which  in  turn  depend  on  the  choice  of  the  measure  of  displacement.     This 


ffc) 
FIG.  2.    Mechanical  System  of  Two  Degrees  of  Freedom  and  Electrical  Equivalent 

choice  is  largely  arbitrary,  and  the  particular  one  here  chosen  may  not  be  the  most  con- 
venient. The  factors  entering  the  problem  which  depend  on  this  arbitrary  choice  are 
termed  generalized.  Thus  s  is  a  generalized  velocity  of  the  system;  in  this  case  it  is  the 
actual  velocity  of  the  mass  M,  but  need  not  be.  On  the  other  hand,  mm  is  not  the  actual 
mass  of  the  system  but  might  be  made  so  with  a  proper  choice  of  s,  the  generalized  ^dis- 
placement. 

30.  ENERGY  OF  MECHANICAL  AND  ELECTRICAL  SYSTEMS 

LINEAR  MOTION.     The  kinetic  energy  of  a  mechanical  system  of  one  degree  of 
freedom  in  linear  motion  is 

.(3) 


where  mm  is  the  generalized  mass  of  the  system  and  s  is  the  generalized  velocity.  The 
generalized  mass  can  be  defined  as  the  quotient  of  the  kinetic  energy  T  of  the  system 
divided  by  the  square  of  the  generalized  velocity. 

The  potential  energy  V  of  the  system  is  a  function  of  the  displacement  of  the  various 
parts  of  the  system;  it  is  independent  of  velocity,  acceleration,  etc.,  and  in  a  system  of 
one  degree  of  freedom,  since  the  displacements  of  the  various  parts  are  all  proportional 
to  the  generalized  displacement  s,  the  potential  energy  V  is  a  function  of  s  only,  that  is 

V  =  V(s)  (4) 

This  can  be  expanded  in  a  Taylor's  series: 


where  VQ  is  the  value  of  V  when  s  =  0,  and  the  subscript  0  indicates  that  derivatives  are 
to  be  evaluated  at  s  =  0.  Since  potential  energy  can  be  measured  from  any  arbitrary 
level,  VQ  may  be  taken  as  zero  without  loss  of  generality.  Furthermore,  in  vibrating  sys- 
tems the  generalized  displacement  may  be  so  chosen  that  s  =  0  for  equilibrium;  then,  since 
dV/ds  =  0  for  equilibrium,  the  second  term  on  the  right-hand  side  of  eq.  (5)  drops  out. 
Making  the  further  assumption  that  terms  containing  higher  powers  of  s  than  the  second  are 
negligible  in  eq.  (5),  V  reduces  to 


where  Cm  is  a  constant  (  —  =  -r-r-      ).    Cm  is  called  the  compliance;  its  reciprocal,  the 
\Cm       oV  Jo/ 

stiffness.    It  will  be  shown  later  that  eq.  (6)  is  equivalent  to  assuming  that  the  restoring 
forces  of  the  system  obey  Hooke's  law. 


5-58  ELECTRIC   CIRCUITS,   LINES,   AND   FIELDS 

If  there  is  dissipation  (heat)  in  the  mechanical  system,  assume  that  the  rate  of  energy 
dissipation  D  is 

D  =  r^s2  (7) 

where  rm  is  a  constant. 

In  many  mechanical  systems,  particularly  those  concerned  with  the  small  motions  usual 
in  acoustic  work,  this  is  practically  true;  it  is  equivalent,  as  shown  below,  to  assuming 
that  the  retarding  frictional  force  is  proportional  to  velocity,  and  for  small  motions  this 
is  approximately  true  of  air  friction.  rm  is  called  mechanical  resistance.  If  it  is  not  a 
constant,  that  is,  if  D  is  not  proportional  to  s2,  but  depends  on  higher  powers  of  s  as  well, 
the  equations  cease  to  be  linear  and  the  methods  of  articles  16-20  must  be  used. 

ROTATIONAL  MOTION.  The  kinetic  energy  of  a  mechanical  system  of  one  degree 
of  freedom  in  rotational  motion  is 

T  =  W#  (8) 

where  I  is  the  moment  of  inertia  of  the  system,  <£  the  angular  displacement  in  radians, 
and  <}>  the  angular  velocity  in  radians  per  second. 
The  potential  energy  is 

"-!! 

where  the  same  assumptions  regarding  higher  powers  of  <j>  are  made  as  in  the  case  of 
linear  motion. 

Likewise  the  dissipation  is  assumed  as 

D  =  rr<p*  (10) 

where  rr  is  a  constant. 

ELECTRIC  CURRENTS.  The  stored  energy  in  an  inductance  L  in  an  electrical 
circuit  is  (cf  .  eq.  [3]) 

T  =  i/aL*  -  i/2L^  (11) 


where  i(=  dq/dt  s  q)  is  the  current  in  the  electric  circuit,  and  q  is  the  charge.    This  is 
often  called  the  kinetic  energy  of  the  electrical  circuit. 
The  energy  in  a  condenser  C  is  (cf  .  eq.  [6]) 

y-£S 

The  rate  of  energy  dissipation  in  an  electrical  circuit  is  (cf  .  eq.  [7]) 

D  =  ri*2  (13) 

where  r  is  the  resistance  of  the  circuit. 

31.  VIBRATIONS  OF  A  SYSTEM  OF  ONE  DEGREE  OF  FREEDOM 

LINEAR  MOTION.  If  a  mechanical  system  of  one  degree  of  freedom  is  in  a  condition 
of  stable  vibration  its  gains  and  losses  of  energy  must  be  equal.  Hence  the  increase  in 
kinetic  and  potential  energy  added  to  the  energy  dissipated  must  equal  the  work  done,  or 

+  «V2Cm  +  frj?  dt  =  fs  (14) 

-where  fs,  the  product  of  force  and  distance,  is  the  work  done  on  the  system.    The  rate  of 
o  change  of  energy  is 

+  rms  +  s/Cm  -  /  (15) 


'Equation  (15)  is  an  equation  in  forces;  that  is,  each  term  has  the  dimensions  of  a  force. 
'The  first  term,  ras,  represents  the  usual  inertia!  force  due  to  the  motion  of  the  system;  the 
:  second  term,  rms,  represents  a  retarding  force,  proportional  to  the  velocity  of  the  system  — 
^this  for  small  displacements  and  velocities  represents  approximately  air  friction,  etc.;  the 
'third  term,  s/Cmi  represents  a  force  within  the  system  proportional  to  the  displacement. 
'This  last  force  is  usually  a  restoring  force,  for  example,  the  restoring  force  of  a  spring. 
I  Since  this  force  is  proportional  to  the  displacement,  Hooke's  law  (that  the  stress  is  equal 

to  the  strain)  applies.  Any  internal  force,  such  as  the  restoring  force  accompanying  the 
;  straining  of  any  part  of  the  mechanical  system,  so  long  as  the  part  is  not  stretched  beyond 

the  elastic  limit,  that  is,  so  long  as  the  restoring  force  is  proportional  to  the  displacement, 

•  contributes  a  term  of  the  form  s/Cm.    On  the  other  hand,  if,  instead  of  a  restoring  force, 

•  the  term  s/Cm  were  to  represent  a  force  proportional  to  the  displacement,  but  tending  not 
i  to  issfcpBe.  the  system  to  its  equilibrium  position  but  to  displace  the  system  further,  then 


COMPARISON  OF  MECHANICAL,   ELECTRICAL  SYSTEMS      5-59 

Cm  would  be  intrinsically  negative.  There  is  no  simple  analog  in  a  passive  electric  circuit 
to  this  negative  Cm  which  sometimes  appears  in  mechanical  systems.  A  "negative  con- 
denser" must  be  used,  such  as  can  be  obtained  in  vacuum-tube  circuits. 

Equation  (15)  would  be  the  equation  of  the  system  of  Fig.  1,  if  the  force  were  applied 
to  the  mass  M  and  if  s  were  measured  from  the  equilibrium  position.  The  force  /  is  a 
function  of  i;  it  can  be  expanded  in  a  Fourier  series  (see  article  2)  and  each  component  of 
Form  F^***  treated  separately.  If  s  =  sj0*  then 


srm  4-  j'wwi™  + 


and  5  =  F/juzm,  where  zm  =  rm  +  jumm  4-  l/jwCm  is  called  the  vector  mechanical  imped- 
ance of  the  mechanical  system.  The  real  part  of  zm  is  the  mechanical  resistance;  the 
imaginary  part  is  the  mechanical  reactance.  If  I  were  determined  instead  of  s,  assuming 
s  =  s^wt,  then  s  =  F/zm.  This  is  identical  in  form  with  the  usual  equation  for  a  sine-  wave 
current  in  an  electric  circuit. 

ROTATIONAL  MOTION.  The  above  analysis  can  be  applied  to  rotational  motion 
if  torque  is  substituted  for  force.  Thus 

rrtp  +  I^  +  ^-L  (17) 

at        LT 

so  that  <}>  =  L/z,  where  z  =  rr  +  /[«/  —  (l/wCr)]  is  called  the  vector  rotational  impedance 
of  the  system. 

32.  COMPARISON   OF  MECHANICAL  AND  ELECTRICAL  SYSTEMS 

UNITS.  The  analogy  developed  above  between  the  equations  of  mechanical  systems 
and  electric  circuits  must  not  be  interpreted  to  mean  that  there  is  any  actual  similarity  or 
analogy  between  quantities  occupying  the  same  position  in  their  respective  equations. 
The  inertia  of  the  electric  circuit  and  the  mass  of  the  mechanical  system  though  appearing 
in  the  same  place  in  the  differential  equation  are  otherwise  quite  diverse.  Perhaps  the 
difference  is  most  convincingly  shown  by  the  fact  that  the  analogous  quantities  current 
and  velocity  have  different  dimensions,  current  having  the  dimensions  M^L^T~l  in  the 
practical  electric  system  whereas  the  dimensions  of  velocity  are  LT~l. 

If  this  difference  is  kept  in  mind  and  it  is  remembered  that  the  analogy  is  merely  a 
formal  one,  representing  the  similarity  of  the  differential  equations  expressing  the  be- 
havior of  the  several  systems,  it  is  convenient  to  draw  analogies  between  all  quantities 
of  the  several  systems.  A  list  of  these  is  given  in  Table  1.  Any  system  of  units  may  be 
used  in  any  of  the  equations;  only  the  cgs  mechanical  systems  and  the  practical  system 
for  the  electric  circuit  are  shown  in  Table  1  on  p.  5-60. 

When  energy  equations  are  written  in  the  two  systems  the  dimensions  of  the  equations 
must  of  course  be  identical,  since  the  dimensions  of  energy  are  fixed.  The  usual  current- 
electromotive  force  equation,  however,  does  not  have  the  same  dimensions  as  a  mechani- 
cal force-velocity  equation.  Nevertheless  it  is  perfectly  proper  to  set  up  electric  circuits 
which  are  equivalent  to  mechanical  systems,  or  vice  versa,  use  the  ordinary  electric  mesh 
equations  to  obtain  a  solution,  and  use  this  solution  to  specify  the  proper  constants  for 
either  or  both  systems.  The  validity  of  this  procedure  depends  on  the  fact  that  the  solu- 
tion of  the  differential  equations  is  independent  of  the  meaning  attached  to  the  symbols. 

SYSTEMS  OF  MANY  DEGREES  OF  FREEDOM.  The  extension  of  the  analogy 
between  electrical  and  mechanical  systems  to  more  than  one  degree  of  freedom,  and  the 
analysis  of  such  systems  which  contain  both  electrical  and  mechanical  portions,  are  most 
readily  accomplished  by  the  application  of  Lagrange's  principle.  In  the  application  of 
this  principle  the  energy  equations  for  the  whole  system  are  first  written  down,  in  terms 
of  measurements  from  an  equilibrium  position.  Lagrange's  equation  that 

d  BT      d(T  -  V)       1  dD  _ 

dft  2  da      Jl  ^     ' 


is  applied  for  each  independent  coordinate  (each  q  needed  in  writing  the  energy  equations). 
ELECTRICAL  CIRCUITS  EQUIVALENT  TO  MECHANICAL  SYSTEMS.  If  the 
energy  equations  of  two  systems  can  be  thrown  into  the  same  form  then  the  solutions 
of  the  two  systems  must  be  identical.  The  same  applies  to  the  differential  equations,  or 
to  the  resulting  mesh  equations,  but  in  general  the  similarity  of  the  systems  can  be  estab- 
lished more  readily  by  use  of  the  energy  equations. 


5-60 


ELECTRIC    CIRCUITS,   LINES,   AND   FIELDS 


Table  1.    Analogous  Quantities  in  Linear  and  Rotational  Mechanical  Systems  and 

Electric  Circuits 

The  cgs  units  and  practical  electrical  units  are  used. 


LINEAB  MOTION 

ROTATIONAL  MOTION 

ELECTRIC  CIRCUIT 

Quantity 

Unit 

Sym- 
bol 

Quantity 

Unit 

Sym- 
bol 

Quantity 

Unit 

Sym- 
bol 

Force 

dyne 

/ 

Torque  

dyne  cm 

L 

Electromo- 
tive force  .  . 

volt 

e 

Displacement 

cm 

s 

Angular 
displacement 

radian 

<f> 

Charge    . 

coulomb 

Q. 

Velocity  

cms 

s 

Angular 
velocity  .... 

radian 

4> 

Current  .... 

ampere 

i 

sec 

sec 

gram 

mm 

Moment  of 
inertia  

gram  cm2 

I 

Inductance.  . 

henry 

L 

Linear  

cms 

cm 

Rotational 
compliance  . 

radians 

Cr 

Capacitance 

farad 

C 

compliance 

dyne 

dyne  cm 

Mechanical 
resistance.  . 

dyne  sec 
cm 
mech.ohm 

rm 

Rotational 
resistance.  .  . 

dyne  cm  sec 

rr 

Resistance  .  . 

ohm 

r 

radian 
rotational  ohm 

Mechanical 
reactance.  . 

ohm 

*m 

Rotational 
reactance.  .  . 

ohm 

xr 

Reactance  .  . 

ohm 

X 

Mechanical 
impedance. 

ohm 

Zm 

Rotational 
impedance.  . 

ohm 

ZT 

Impedance 

ohm 

z 

p 

ergs 
sec 

Pm 

Power  .    .  •  • 

ergs 
sec 

Pr 

watts 

P 

The  rules  for  establishing  the  equivalence  of  an  electric  circuit  with  a  mechanical 
system  are  therefore: 

1.  Write  the  equations  for  the  kinetic  energy,  the  potential  energy,  and  the  dissipation 
of  the  mechanical  system. 

2.  There  will  be  an  electric  mesh  for  each  independent  generalized  coordinate. 

3.  Any  parameter  which  appears  in  an  energy  equation  multiplied  only  by  the  square 
of  one  generalized  coordinate  is  a  part  of  the  corresponding  electric  mesh,  and  is  not 
common  to  any  other  mesh. 

4.  Any  parameter  which  appears  multiplied  by  the  difference  of  two  generalized  coordi- 
nates will  be  common  to  the  corresponding  electric  meshes. 

5.  Any  parameter  which  appears  multiplied  by  the  product  of  two  generalized  velocities 
may  be  represented  by  a  mutual  inductance  between  the  corresponding  electric  meshes. 

6.  If  parameters  appear  in  the  energy  equations  multiplied  by  any  other  combinations 
of  the  generalized  coordinates,  new  coordinates  should  be  chosen  in  an  attempt  to  eliminate 
them.    If  this  is  impossible  *  there  is  no  one-to-one  (parameter-to-parameter)  equivalent 
circuit. 

To  illustrate  the  method  two  examples  will  be  worked  out. 

In  the  mechanical  system  shown  in  Fig.  2  (a)  assume  that  the  masses  MI  and  M%  are 
resting  on  a  plane  surface  whose  frictional  force  is  proportional  to  velocity.  Let  Si  and  S2 
be  the  changes  in  s'i  and  s'a  from  the  equilibrium  position.  Then 


T  = 


S2)2 


and 


2  Cm 


*  Such  cases  will  not  be  frequent.  When  encountered  it  is  best  to  write  force  equations  and  attempt 
to  obtain  circuits  which  will  represent  the  mechanical  system  by  allowing  electrical  parameters  to 
represent  combinations  of  mechanical  constants;  for  example,  Z/2  =  2m2  +  mi  or  C$  =  Cm\  —  Cm4 
-f-  Cms  might  be  required.  In  many  such  cases  the  utility  of  the  method  is  doubtful. 


COMPARISON   OF  MECHANICAL,   ELECTRICAL  SYSTEMS      5-61 


The  equations  for  T  and  D  both  contain  a  parameter  multiplied  by  the  sum  of  two  gen- 
eralized velocities  squared,  for  which  no  electrical  equivalent  is  given  in  the  usual  simple 
rules  for  independent  meshes.  These  equations  may  be  altered  in  two  ways;  first,  the 
direction  in  which  s'2  (or  s'i)  is  measured  may  be  changed;  or  second,  new  s's  may  be  chosen. 
If  the  first  alternative  is  employed 


V  is  unchanged  and 


T  - 
D  = 


The  equivalent  electrical  circuit  is  shown,  in  which  LI  —  MI,  L2  =  M^  r\  =  rmi,  r%  =  rm2l 
d  SB  Cmi  and  Cz  =  Cm2.  That  is,  in  the  electrical  circuit  1  henry  of  inductance  is  specified 
for  each  gram  of  mass  in  the  mechanical  system,  1  farad  of  capacitance  for  each  centi- 
meter/dyne of  compliance,  etc. 


V       2  CW 


Cm2 


FIG.  3.    Same  Mechanical  System  as  Fig.  2  and  Alternate  Electrical  Equivalent 

If  the  second  alternative  is  chosen  the  sketch  of  Fig.  3  would  represent  the  conditions. 
Here 

1  (s2  —  si)2 
'2" 
and 

D  = 
The  corresponding  electric  circuit  is  shown. 

It  is  thus  possible  to  have  more  than  one  equivalent  electric  circuit  for  a  given  mechan- 
ical system,  depending  on  the  choice  of  independent  variables  for  the  mechanical  system. 
Casual  examination  of  the  electric  circuits  reveals  them  as  equivalent,  in  that  although 
the  mesh  currents  differ  the  current  through  any  element  would  be  the  same. 


FIG.  4.    Mechanical  System  of  Four  Degrees  of  Freedom  and  Electrical  Low-pass  Filter 

In  the  mechanical  system  shown  in  Fig.  4  first  assume  that  the  displacements  are 
measured  between  masses,  as  in  Fig.  2.    The  energy  equations  are 

T  = 


2 


and 


In  this  case  it  is  impossible  to  throw  the  equations  into  the  usual  mesh  equation  form  by 
changing  the  direction  of  measurement  of  one  or  more  of  the  variables,  so  the  variables 
must  be  changed  if  the  convenience  of  the  mesh  equation  technique  is  to  be  utilized.  If 
the  system  used  in  Fig.  3  is  assumed 

T  -  lyWi^i2  +  i/2^22  +  VaAfjA2  + 


2 


-  Si)2 


and 


The  equivalent  electric  circuit  is  a  low-pass  filter  as  shown. 


5-62 


ELECTKIC   CIRCUITS,  LINES,   AND   FIELDS 


33.  ELECTROMECHANICAL-ACOUSTIC  SYSTEMS 

Many  vibrating  systems  consist  of  combinations  of  two  or  more  of  the  separate  energy 
systems  discussed  above.  Examples  of  such  combinations  are  electrically  controlled  vi- 
brating reeds  and  tuning  forks,  microphones  and  speakers  of  all  varieties,  fluid-type  auto- 
mobile stabilizers,  and  almost  all  musical  instruments. 

In  such  systems  energy  is  converted  from  one  form  to  another  within  the  system.  The 
methods  of  converting  energy  are  many,  so  that  the  forms  of  the  interaction  between  the 
different  portions  of  the  system  are  many,  but  the  same  general  method  of  deterrnining 
the  force  equations  that  was  used  for  homogeneous  systems  can  be  applied. 

The  energy  of  each  portion  of  the  system  should  be  set  up  exactly  as  for  a  homogeneous 
system  in  terms  of  the  constants  of  the  particular  portion.  Some  of  these  constants  will 
be  functions  of  the  velocity  or  displacement,  etc.,  of  some  other  portion  of  the  system, 
and  these  must  be  so  specified.  Then  the  "force"  equations  may  be  obtained  directly  by 
means  of  Lagrange's  equation,  and  from  these  the  mesh  equations  may  be  written.  These 
mesh  equations  will  not  usually  be  of  the  same  dimensions,  but  this  fact  may  be  disregarded 
since  the  interaction  factors  between  the  equations  will  be  such  that  when  the  equations 
are  solved  simultaneously  these  differences  will  be  automatically  compensated.  One  set 
of  self-consistent  units  is  shown  in  Table  2.  Several  particular  cases  will  be  developed 
to  show  the  operation  of  the  method. 

ELECTROMECHANICAL  SYSTEMS,  ELECTROSTATICALLY  COUPLED.  Con- 
sider a  system  consisting  of  two  metal  plates,  one  fixed  and  immovable,  the  other  held  in 
position  by  a  spring  (cf.  Fig.  5)  and  resting  on  a  rough  surface.  An  electric  circuit  con- 
nected between  the  two  plates  contains  an  inductance,  a  resistance,  and  a  source  of  voltage. 


Equilibrium 


FIG.  5.    Electromechanical  System,  Electrostatically  Coupled 

The  potential  energy  Vm  stored  in  the  spring  (measuring  energy  from  its  value  at  the 
equilibrium  position,  when  the  electric  charge  is  zero)  is 

7    =!-* 

m       2Cm 

where  Cm  is  the  compliance  of  the  spring  and  $2  is  the  distance  the  spring  is  stretched  by 
moving  the  movable  plate  toward  the  fixed  plate. 
The  potential  energy  Ve  stored  in  the  condenser  is 


where  q  is  the  charge  on  the  condenser  and  C  its  capacitance.    The  value  of  C  is 

c_         kS         =      C' 

47r(s0  -  s2)        s0  —  $2 

where  k  is  the  dielectric  constant,  S  the  cross-section  of  the  plates,  and  (so  —  s)  the  distance 
between  them. 

The  kinetic  energy  of  the  system  is 

m*l      Lf 
2^2 

where  m  is  the  generalized  mass  of  the  movable  plate  and  spring  together  and  L  is  the 
inductance  of  the  electrical  system.    The  dissipation  of  the  system  is 

D  =  rJ2       ri? 


ELECTROMECHANICAL-ACOUSTIC  SYSTEMS 


5-63 


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5-64 


ELECTRIC   CIRCUITS^   LINES,   AND  FIELDS 


The  force  equations  may  be  obtained  by  applying  Lagrange's  equations  to  these  energy 
equations;  they  are 

so          <r 
m$2  +  ~^r  —  ST£  4-  rmsz  =  / 

Um         4L> 

and 


Assuming  that  q  can  be  split  into  a  steady  or  average  part  g0  and  a  varying  part  qi  and 
that  qi  is  considerably  smaller  than  g0,  so  that  squares  of  qi  are  negligible, 


_  _ 

20'  2C" 

The  first  term  on  the  right-hand  side  of  this  equation  represents  a  steady  pull  or  initial 
set  of  the  plunger  accompanying  the  initial  charge  on  the  condenser.     The  second  term 

reduces  to  —  qi,  and  the  third  term  is  negligible  by  assumption.    Neglecting  the  steady 
c/ 

terms  the  force  equations  become 


+  rmsz  =  fz 


Qo 

' 


(19o) 

(196) 

U.fc  \s  O 

which  for  steady-state  sine  waves  may  be  written  in  the  familiar  mesh  equation  form 

FZ   =    222*2  4"  2i2/l  (20a) 

El   =   2i252  4-  2n/l  (206) 

where  mesh  2  represents  the  mechanical  and  mesh  1  the  electrical  system,  zu  and  222 
are  the  usual  mechanical  and  electrical  impedances  and  Fz  and  EI  the  respective  "forces." 
z12  =  —  qo/juC'  =  ~  l/7*cdCi2  is  the  interaction  factor.  The  dimensions  of  I/ CM  are  such 
that  multiplication  of  it  by  a  charge  gives  a  quantity  with  the  dimensions  of  a  force, 
while  multiplication  of  it  by  a  distance  or  length  produces  a  quantity  with  the  dimensions 
of  an  electromotive  force.*  It  is  energy  divided  by  distance  and  charge,  and  its  use 
is  valid  for  any  system  of  electrical  units  in  combination  with  any  system  of  mechanical 
units  provided  those  units  were  used  in  evaluating  ZM. 

Equations  (20)  are  then  in  proper  form  to  use  for  any  case  of  mechanical  and  electrical 
systems  where  the  interaction  between  the  two  systems  depends  on  varying  the  capacitance 
of  the  electrical  circuit  by  the  motion  of  a  part  of  the  mechanical  system. 

ELECTROMECHANICAL  SYSTEMS,  MAGNETICALLY  COUPLED.  When  the 
interaction  between  the  electric  and  mechanical  systems  occurs  through  magnetic  attrac- 


FIG.  6.    Electromechanical  System,  Electromagnetically  Coupled 

tion  the  inductance  of  the  electric  circuit  is  varied  by  the  mechanical  motion.    The  energy 
equations  for  the  system  of  Fig.  6  are  (s0  is  the  equilibrium  position) 


and 


V  =  ~ 
T  =  —c 

D  =  rmJ22  4- 


L 


*  The  factor  l/Ci2  has  the  dimensions  of  an  electric  field,  although  it  is  a  rather  unusual  expression 
for  such  a  field. 


ELECTROMECHANICAL-ACOUSTIC  SYSTEMS 


5-65 


Here  L,  the  self-inductance  of  the  electric  circuit,  is  a  function  of  the  displacement  s^. 
Since  most  of  the  reluctance  of  the  magnetic  circuit  is  in  the  air  gap  rather  than  in  the 
iron  portions,  L  can  be  assumed  to  vary  as  the  first  power  of  s  for  small  displacements,  with 
little  error.  Hence 

L  =  L0(l  +  6s2) 

where  b  is  a  constant  depending  on  the  configuration  of  the  physical  system  and  on  the 
systems  of  units  used. 

Applying  Lagrange's  equations,  the  force  equations  for  the  system  are 


+  eT 


sz  -  f 


and 


L  -- 


Assuming  that  i  is  composed  of  a  steady,  or  average,  value  ZQ  and  a  varying  part  ii,  and 
that  i\  is  considerably  smaller  than  t"0,  so  that  squares  of  i\  are  negligible, 


The  first  term  on  the  right-hand  side  again  represents  a  steady  pull  or  initial  set  of  the 
plunger  accompanying  the  average  or  d-c  exciting  current  of  the  magnet.  The  second 
term  reduces  to  —  (&Loio)ii,  and  the  third  term  is  negligible  by  assumption.  Disregarding 
the  steady  terms  the  force  equations  become 


and 


which  for  steady-state  sine  waves  may  be  written 
EI  =  zuli  +  zi2S2 


(2  la) 
(216) 

(22a) 

^2   =   Zn.Ii  +  Z22S2   =    —   212/1   +  Z22$2  (226) 

where  mesh  1  represents  the  electrical  and  mesh  2  the  mechanical  system,  EI  and  F% 
represent  the  respective  "forces."  In  this  case,  however,  the  interaction  factors  are  not 
equal,  but  212  =  —  221. 

The  solutions  for  current  and  velocity  give 


/     _ 


and 


indicating  that,  when  a  mechanical  force  is  applied  to  the  plunger  in  phase  with  an  applied 

electromotive  force,  the  velocity  of  the  plunger  is  increased  but  the  electric  current  is 

decreased.     Analysis  of  the  physical  action  also  indicates 

this  result  since  increase  in  the  electromotive  force  attracts 

the  plunger,  but  the  inward  motion  of  the  plunger  due  to  an 

applied  force  increases  the  inductance  of  the  electric  circuit 

and  so  decreases  the  current  flow. 

These  are  the  equations  of  a  simple  moving-armature 
telephone  receiver  such  as  that  shown  in  Fig.  7,  when  eddy 
currents  are  neglected.  The  input  impedance  (zt-)  of  the 

receiver  is  the  ratio  of  EI  to  /i,  when  there  is  no  externally  

applied  force  (that  is,  when  F%  =  0) ,  hence  p 


_          , 

—  Zn  -f 


FIG.  7.    Simple  Telephone  Re- 
ceiver  (No  Eddy  Currents) 


The  blocked  impedance  of  a  receiver  is  defined  as  the  impedance  measured  when  the 
diaphragm  is  constrained  from  moving  (diaphragm  held  at  s  =  0)  .  The  motional  imped- 
ance is  the  difference  between  Zi  and  the  damped  impedance;  that  is,  it  is  that  part  of  z* 
due  to  the  motion  of  the  diaphragm.  Evidently  for  the  simple  receiver  the  damped  im- 


5-66  ELECTRIC  CIRCUITS,   LINES,   AND  FIELDS 

pedance  is  z\\  (which  is  the  impedance  of  the  electric  circuit  considered  alone)  and  the 
motional  impedance  2mot.  is 

2mot.  =  —  (23) 

222 

Since  zia  is  a  constant,  independent  of  frequency  by  definition,  it  follows  that,  if  the  real 
part  of  Zmot.  (the  resistance  component  of  zmot.)  and  corresponding  values  of  the  imaginary 
component  (reactance  component)  be  plotted  for  various  frequencies,  the  resulting  graph 
will  be  a  circle,  called  the  motional  impedance  circle,  tangent  to  the  j  axis,  with  center  on 
the  positive  real  axis  (see  High-frequency  Alternating  Currents,  Mcllwain  and  Brainerd, 
John  Wiley  &  Sons,  Chapter  XIII). 

MECHANICAL-ACOUSTIC  SYSTEM.     The  simplest  mechanical-fluid  system  is  that 
where  a  plunger  moves  in  a  fluid  displacing  a  volume  of  the  fluid  (see  Fig.  8) . 

^^^^^^^^^_^        The  potential  energy  of  a  volume  of  fluid 
^  t%#£#%^^  moving  as  a  unit  is 

%?  "jvtf 

ffl  ^"m.  '^R$|  ,          lUf  . 

Eluid  Tf  =  l/zplSsP  =  — -  S2 

>.,  Qp  mf        where  p  is  the  density,  I  the  length,  S  the 
^^    cross-section,  and  m/  the  mass  of  the  fluid,  and 


, 

FIG.  8.    Simple  Mechanical-acoustic  System      s  is  the  particle  velocity.     If  the  fluid  is  in- 

compressible, s  for  the  fluid  will  equal  I,  the 

velocity  of  the  plunger,  and  mf  will  be  the  actual  mass  of  the  fluid;  if  the  fluid  is  compressi- 
ble I  may  still  be  taken  as  the  velocity  of  the  plunger  but  the  generalized  mass  of  the  fluid 
will  be  less  than  the  actual  mass.  The  potential  energy  and  dissipation  are 


and 

Df  =  r/ 

The  total  energy  equations  of  the  system  are 

2Cf 

^  J 

- 

and 

D  =  rms?  +  r/s2 

where  r/  and  C/  as  well  as  m/  are  generalized  parameters.     Operating  with  Lagrange's 
equation  there  results 

T-  +  -r)  *  +  (mm  +  m/)s  +  (rm  +  r/)i  (24) 

Thus  this  combination  of  a  mechanical  and  an  acoustic  system  can  be  represented  by  one 
equation  with  only  one  coordinate,  instead  of  the  two  equations  which  might  have  been 
expected.    Each  parameter  of  the  mechanical  system  has  been  altered  by  the  presence 
of  the  acoustic  chamber,  and  the  resonance  frequency  has  been  lowered. 
Assuming  sine-wave  excitation,  eq.  (24)  becomes 

Fm  -  Ff  =  \  (rm  +  rf)  +  j  (wmm  +  umf  -  —  --  *>Cf  J    I 
or 

Fm  ~   Ff  =  ZmfS 

where 


and 


umm  +  com/ 


8mf  =  tan-' 


—  - 

CfC 


The  net  result  of  the  air  chamber  is  thus  to  increase  the  resistance  and  inertial  effect 
of  the  plunger  but  to  decrease  its  compliance.  In  general  both  the  magnitude  and  phase 
angle  of  the  impedance  are  altered. 


BIBLIOGRAPHY  5-67 

BIBLIOGRAPHY 

Mcllwain,  K.,  and  J.  G.  Brainerd,  High-frequency  Alternating  Currents.    John  Wiley  (1939). 
Olson,  H.  P.,  and  F.  Massa,  Applied  Acoustics.    Blakiston  (1934). 


_       (1927). 
L  (1928). 
i  Nostrand  (1942). 


Articles 


Wegal,  R.  L.,  Theory  of  Telephone  Receivers,  J.  AJ.E.E.,  Vol.  40,  791  (October  1921). 

Hussey,  L.  W.,  and  L.  R.  Wrathall,  Oscillations  in  an  Electromechanical  System,  B.S.T.J.,  VoL  15, 

441  (July  1936). 
Mason,  W.  P.,  and  R.  A.  Sykes,  Electrical  Wave  Filters  Employing  Crystals  with  Normal  and  Divided 

Electrodes,  B.S.T.J.,  Vol.  19  (April  1940). 
Mason,  W.  P.,  Electrical  and  Mechanical  Analogies,  B.S.T.J.,  Vol.  20,  405  (October  1941). 


SECTION  6 
PASSIVE  CIRCUIT  ELEMENTS 


SINGLE-MESH    AND    COUPLED    CIRCUITS 

BY  VEKNON  D.  LAND  ON  AND 

ART.  K*0*    MdLWAIN  pAQE 

1.  Series  Resonant  Circuits 02 

2.  Parallel  Resonant  Circuits 04 

3.  Attenuators,  Pads 05 

4.  Coupled  Circuits 06 

5.  Currents  and  Voltages  in  Coupled  Cir- 

cuits   07 

6.  Air-core  Transformers 10 

7.  Three-winding     Transformers     (Hybrid 

Coils) 12 

TRANSFORMERS  WITH  IRON  CORES 
BT  A.  J.  ROHNER 

8.  Audio-frequency  Transformers 13 

9.  Output  Transformers 17 

10.  Input  and  Interstage  Transformers 19 

11.  Driver  Transformer 22 

12.  Physical  Design  of  Audio  Transformers  22 

13.  Audio  Transformer  Measurements 25 

14.  Power  Transformer 26 

15.  Vibrator  Transformer 30 

16.  Pulse  Transformer 32 


ELECTRIC  WAVE  FILTERS 

^•r,  BT  A.  J.  GHOSSMAN  PAGE 

17.  Introduction 33 

18.  Properties  of  the  Image  Parameters.  ...  36 

19.  Open-circuit  Transfer  Impedance 38 

20.  Transfer  Constant  Theorem 39 

21.  Image  Impedance  Theorem 39 

22.  The  General  Composite  Filter 40 

23.  Symmetrical  Sections 41 

24.  Unsymmetrical  Sections 50 

25.  Tchebycheff  Type  Characteristics 56 

26.  Mayer's  Theorem 61 

RADIO  ANTENNAS 

BT  J.    C.   SCHELLENG 

27.  Principles  of  Linear  Conductor  Antennas  65 

28.  Principles  of  Directivity 71 

29.  Directivity    of    Linear    Conductor    An- 

tennas    73 

30.  Directivity    of    Quasi-optical    Antennas 

and  Horns 76 

31.  Practical  Antenna  Systems 80 

32.  Direction  Finding 87 

33.  Miscellaneous 88 


6-01 


PASSIVE  CIRCUIT  ELEMENTS 
SINGLE-MESH  AND  COUPLED  CIRCUITS 

By  Vernon  D.  Landon  and  Knox  Mcllwain 

The  individual  elements  used  in  communication  circuits  must  be  carefully  designed  to 
meet  certain  definite  specifications  if  the  circuit  as  a  whole  is  to  function  properly.  Among 
the  more  important  factors  to  be  considered  are  the  efficiency  and  load-carrying  capacity 
of  the  element,  its  impedance,  its  selectivity,  and  the  introduction  of  the  various  forms  of 
distortion. 

Circuit  requirements  frequently  demand  not  only  that  the  transmission  (ratio  of  output 
to  input)  have  a  particular  value  at  a  given  frequency,  but  also  that  the  variation  of 
transmission  with  frequency,  called  the  transmission-frequency  characteristic,  have  a 
definite  form. 

1.  SERIES  RESONANT  CIRCUITS 

Figure  1  represents  a  series  circuit  in  which  the  resistance  of  all  the  elements  of  the  cir- 
cuit is  lumped  into  one  resistance  r,  and  similarly  for  the  inductance  and  capacitance.  For 
a  given  value  of  impressed  voltage  the  current  in  this  circuit  depends  on  r,  L,  and  C,  and 

the  frequency  of  the  impressed  voltage.  For  given 
values  of  L  and  C  there  will  be  one  frequency,  and 
only  one,  for  which  coL  =  1/coC.  At  this  frequency 
(fr},  which  is  called  the  resonant  frequency  of  the 
circuit,  the  current  is  maximum  for  any  given  value 
of  voltage  and  is  in  phase  with  the  impressed  volt- 
age. The  resonant  frequency  is  given  by 


FIG.  1.    Simple  Series  Resonant  Circuit 


fr- 


(1) 


where  L  is  in  henrys  and  C  in  farads.    When  L  is  expressed  in  millihenry s  and  C  in  micro- 
farads eq.  (1)  becomes: 

5033  .,   . 

'' "  VTc  (U) 

The  absolute  value  of  the  current  Ir  in  the  circuit  at  this  frequency  is 


(2) 

For  frequencies  less  than  fr,  coL  <  — — ;  and  the  current  will  lead  the  impressed  voltage 

(/3  is  negative),  whereas  for  frequencies  greater  than  fr  the  current  will  lag  the  voltage 
(j3  is  positive,  see  Section  5,  article  2) . 

In  Fig.  2  are  shown  graphs  of  the  absolute  values  of  current  versus  frequency  in  a 
simple  series  circuit  for  two  values  of  resistance. 

VOLTAGE  RELATIONS.  At  the  resonant  frequency  the  magnitude  of  the  impressed 
voltage  is  given  by  E  =  rl,  and  the  voltages  across  the  inductor  and  condenser  are 

2irfrLI  and  Ec  —  0_r^  .    Since  the  only  condition  necessary  for  resonance  is  that 


the  sum  of  the  reactances  is  zero,  either  reactance  may  be  many  times  as  great  as  the 
circuit  resistance;  thus  the  voltage  across  the  inductor,  or  the  condenser,  may  be  several 
hundred  times  the  impressed  voltage.  Figure  3  shows  how  the  voltage  components  in  a 
circuit  vary  as  the  frequency  is  changed. 

6-02 


SERIES  RESONANT  CIRCUITS 


6-03 


24 
22 
20 
18 
£16 

a 

Jl4 
k 

J>12 

"c 

£ 

3  10 

8 
6 
4 
2 
0 

\ 



1 

L~ 

655  Mh. 
0.0605  Mf. 
60  Ohms 
1060  Ohms 
1.3  Volts 

rz- 

?*•— 

E= 

J 

L-i 

i 

1^ 

\ 

5 

o-? 

J 

\ 

,E 

-  = 

Q. 
< 

V 

p 

-**&* 

•-"*"" 

i?> 

-^> 

f*-^ 

400              500           600      700       800             1000        1200     1400   1600 
Frequency^  Cycles  per  Second 

2<X 

FIG.  2.     Variation  of  Current  with  Frequency  in  a  Simple  Series  Circuit  (Ji  is  for  resistance  n,  and 

/2  for  rs) 


400 


500 


600      700      800  1000 

Frequency  in  Cycles  per  Second 


1200     1400    1600 


2000 


FIG.  3.    Variation  of  Voltage  Components  with  Frequency  in  a  Simple  Series  Circuit 


6-04 


PASSIVE   CIRCUIT  ELEMENTS 


CIRCUIT  Q.    The  per  cent  change  in  current  from  its  maximum  value  for  a  given 
percentage  frequency  change  depends  on  the  relative  values  of  r,  L,  and  C.    As  a  measure 

of  this  change  a  factor  Q  is  defined  as  oxL/r. 
From  this  the  voltage  across  the  coil  (or 
condenser)  of  a  simple  series  circuit  at  re- 


0.7 


3  0.6 


;  0.4 


'0.3 


7 


\/ 


60 


40 


20 


-40 


-60 


-80 


-2.0  -1.5  -1.0  -0.5      0      0.5 
Value  of  a 


1.0    1.5    2.0 


FIG.  4.     Values  of  a  Universal  Resonance  Curve. 
Calculated  from 


1+  ~  +  3  a 


sonance  is 


2irfrLE 


=  EQ 


(3) 


For  a  given  value  of  inductance  the  circuit 
having  the  higher  Q  will  have  smaller  r,  will 
have  higher  resonance  current  in  compari- 
son to  current  off  resonance,  will  be  repre- 
sented by  a  more  sharply  peaked  current 
vs.  frequency  curve,  and  is  said  to  be  more 
sharply  tuned. 

Universal  Resonance  Curve.  Terman 
has  shown  that,  when  Q  is  constant  (loss 
resistance  proportional  to  frequency),  the 
relation  of  resonant  current  to  current  off 
resonance  for  any  circuit  can  be  plotted  in  a 
universal  resonance  curve  as  shown  in  Fig.  4. 
If  the  ratio  of  the  actual  current  and  the 
current  at  resonance  are  known,  the  ordi- 
nate  is  established  and  therefore  the  two 
possible  values  of  a.  From  these  the  re- 
quired deviation  in  frequency  is  determined 
from  the  relation 


-  Q 


Cycles  off  resonance 
Resonant  frequency 


(4) 


Conversely,  if  the  deviation  in  frequency  is 


since  a/Q  is  quite  small.     For  paraUel  circuits    ^nversely,  it  the  deviation  in  frequency  is 
change  ratio  of  currents  to  ratios  of  impedances    •lmown>  a  can  be  readily  found  and  thence 

ft,r\f{    Ifo.Hl'nO1  'nhn.SR  fl.nfrlps  f.n  laororincr  +.ll»    Vola+.-lTrii    Tac-r\r\r\  DQ 


o     mp 
and  leading  phase  angles  to  lagging. 


•> 

the  relative  response. 


2.  PARALLEL  RESONANT  CIRCUITS 

It  is  not  so  easy  to  define  resonance  in  a  parallel  circuit.     If  the  condition  xi  =  wL  = 
-  xz  =  l/o>C  is  used,  there  results 


and 


n-hs 


(5) 


x  =  ri  +r2  Xl  (6) 

so  that  as  n  and  rz  decrease  r  rises  indefinitely  and  theoretically  becomes  infinite  when 
the  circuit  resistance  is  zero.  However,  x  is  not  zero  at  the  fre- 
quency/,., so  that  the  current  is  not  in  phase  with  the  voltage,  nor 
is  s  a  maximum.  Either  of  these  last  conditions  may  be  the 
important  one,  and  the  condition  of  z  maximum  (I  minimum)  is 
certainly  the  easiest  to  measure.  If  n  —  ro  the  three  conditions 
coincide,  and  are  but  slightly  different  when  n  and  rz  are  small. 
In  the  particular  important  case  of  a  coil  and  condenser  in 
parallel  (Fig.  5)  these  equations  reduce  (neglecting  the  resistance 
of  the  condenser)  to 


FIG.  5.     Simple  Parallel 
Resonant  Circuit 


r  =  — -     and 
rLC 


and  the  frequency  for  which  x  =  0  is 


(£-• 


(7) 


(8) 


Graphs  of  r,  x,  and  Z  for  such  a  circuit  are  shown  in  Fig.  6. 


ATTENUATORS,  PADS 


6-05 


200,000 


160,000 


£120,000 


2    80,000 


40,000  - 


700  800  900 

Frequency  in  Cycles  per  Second 


1100      1200 


•f  60,000 


+40,000 


-h  20,000 


-20,000 


-40,000 


—60.000 


FIG.  6.    Variation  of  the  Impedance  of  a  Simple  Parallel  Circuit  with  Frequency 

Another  case  of  particular  interest  exists  when  TL  —  TC  =  VL/C.  Then  for  any 
frequency 

r  -  VL/C  =  TL  =  re    and    x  =  0  (9) 

Thus  the  apparent  resistance  is  constant  with  frequency,  and  the  apparent  reactance  is 
zero  for  all  frequencies. 

TUNING.  Although  they  are  subject  to  the  same  limitations  as  regards  controlling 
selectivity  and  frequency  characteristic  as  the  series  resonant  meshes,  simple  parallel 
tuned  circuits  are  frequently  used  as  coupling  elements  between  vacuum  tubes.  When 
the  circuit  of  Fig.  5  is  adjusted  for  unity  power  factor,  Ie  =  EXL/(T*  +  XLZ),  IL  = 
.E/Vr2  +  XLZ,  Iz  —  ErC/L,  and  z  =  L/rC.  If  the  capacitance  only  is  varied,  unity  power 
factor  coincides  with  the  condition  of  Tnii-iin-mm  line  current  and  with  maximum  impedance. 

When  the  circuit  is  tuned  by  adjustment  of  the  inductance  this  is  not  true,  and  in  this 
case  full  output  is  obtained  for  a  slight  detuning  from  the  condition  of  minimum  plate 
current.  If  the  resistance  remains  constant  the  maximum  impedance  is 


where 


XL  =  r 


-  1 


whereas  if  the  phase  angle  (XL/T)  of  the  inductance  remains  constant  the  maximum  im- 
pedance is  z  =  vr2  -+-  #L2  XL/T  which  occurs  when  XL  =  xc- 
(For  further  details  see  R,.  Lee,  Proc.  I.R.E.,  Vol.  21,  271.) 


3.  ATTENUATORS,  PADS 

Use  of  the  concepts  and  equations  of  insertion  loss  (Section  5,  article  7)  permits  the 
designs  of  definite  elements  (either  dipoles  or  quadripoles)  which  may  be  inserted  into 
circuits  to  produce  definite  effects  at  a  given  frequency.  Such  elements  are  called  attenu- 
ator sections,  or  sometimes  pads;  more  complicated  sections  which  aim  to  provide  a  desired 
definite  variation  of  loss  with  frequency  are  termed  distortion  correctors,  or  equalizers. 


6-06 


PASSIVE   CIRCUIT  ELEMENTS 


MATCHED  IMPEDANCES.  The  simplest  design  occurs  when  it  is  desirable  to  have- 
uniform  loss  at  all  frequencies,  in  which  case  the  image  impedances  of  the  network  must 
match  those  of  the  circuit  into  which  it  is  inserted.  By  Thevenin's  theorem  (p.  5-12)  the 
circuit,  no  matter  how  complicated,  can  be  reduced  to  a  generator  in  series  with  a  simple 
impedance  on  each  side  of  the  point  of  insertion.  (See  Figs.  7  and  8.) 

These  impedances  are  designated  by  Z  and  z  and  their  ratio  by  Z/z  =  s2.  Either  T  or 
TT  type  sections  may  be  utilized.  The  formulas  are  most  useful  for  resistive  networks. 


FIG.  7.    T  Section  Attenuator 

T  SECTIONS.  To  design  a  T  section  (see  Fig.  7)  which  causes  a  loss  of  n  decibels,  read 
off  the  current  ratio  (k)  corresponding  to  such  a  loss  in  the  decibel  table,  Section  1.  Then 
the  proper  values  for  the  arms  of  the  T  section  are 


-  2k/  s 


I  +  k*  -  2ks\ 

i-g  r  and 


k 


When  the  impedances  in  either  direction  are  equal, 

1  -  k 


1  —  z,     then     s  =  1     and    u  =  v  •• 


1  +  k 


z     and    w 


2k 


For  instance,  to  cause  a  loss  of  3  db  in  a  circuit  where  Z  =  z  =  500  /  60°,  k  =  0.7080, 
u  =  v  =  85.5  Z60°,  and  w  =  1420  Z60°. 

To  design  a  balanced  T  or  H  section  simply  put  */2  u  and  1/2  v  in  each  series  leg. 

TT  SECTIONS.     The  proper  values  for  the  arms  of  the  TT  section  of  Fig.  8  are 


-  A? 


sz  _ 

'  and 


-  kz 


When  the  impedances  in  either  direction  are  equal,  Z  =  z,  then  s  =  1  and  u  ~  w 


FIG.  8.    v  Section  Attenuator 

Tabular  aids  in  computing  such  sections  and  rules  for  other  types  of  sections  are  given, 
by  McElroy  (Proc.  I.R.E.,  Vol.  23,  213  [March  1935]). 

NON-MATCHED  IMPEDANCES.  When  the  impedance  variation  with  frequency 
of  the  terminating  circuits  is  such  that  it  cannot  be  duplicated  in  a  simple  network,  the 
insertion  loss  of  the  network  at  any  frequency  can  be  obtained  by  the  methods  of  Section 
5,  article  8.  Exact  methods  of  design  of  sections  to  insure  a  predetermined  variation  of 
loss  (and  phase  change)  with  frequency  are  given  by  Mead  (B.S.T.J.,  Vol.  7,  195), 
Zobel  (B.S.T.J.,  Vol.  7,  438)  and  Gewertz  (Network  Synthesis,  Williams  and  Wilkins 
Co.  [1933]). 

4.  COUPLED  CIRCUITS 

Two  electric  meshes  are  said  to  be  coupled  to  each  other  when  they  have  an  impedance 
in  common,  so  that  a  current  in  one  causes  a  voltage  in  the  other.  The  common,  or 
mutual,  impedance  is  defined  as  the  factor  by  which  the  current  in  one  mesh  must  be 


CURKENTS  AND  VOLTAGES  IN  COUPLED   CIRCUITS      6-07 

multiplied  to  give  the  voltage,  due  to  that  current,  in  the  second  mesh.  It  may  be  a  pure 
resistance,  or  a  pure  reactance,  in  which  case  the  meshes  are  said  to  have  a  pure  coupling- 
or  the  common  impedance  may  be  complex,  in  which  case  the  coupling  is  said  to  be  complex 
Figures  9  and  10  illustrate  some  of  the  more  usual  types  of  coupling,  although  a  com- 
plicated intervening  network  may  be  considered  as  simply  a  coupling  impedance  if  the 
relation  between  current  in  one  mesh  and  resulting  voltage  in  the  other  is  of  interest 


Mesh  a  <      Mesh  6  Mesh  a    §        Mesh  6 


A.  Resistance  Coupling  B.  Inductance  Coupling 


Mesh 


Mesh  6 


C.  Mutual  Inductance 
Coupling 


D.  Capacitance  Coupfing 


FIG.  9.    Simple  Types  of  Coupling 

MUTUAL  IMPEDANCES.  The  mutual  impedances  of  the  circuits  shown  in  Fig.  9 
are  ZM  =  r,  juL,  jcoM,  1/jcaC.  In  Fig.  10,  case  E,  the  mutual  impedance  is  ju(L'  +  Jlf) 
if  the  windings  are  in  the  same  sense  and  jut(L'  —  M)  if  they  are  in  opposite  sense.  In 
case  F,  ZM  =  jfaL  -  1/coC);  in  case  G,  ZM  =  jfaL/oPLC  -  1).  When  the  middle  mesh 
of  case  H  is  considered  as  a  mutual  impedance  between  meshes  a  and  b  its  value  is 


when  the  transformer  windings  are  so  connected  as  to  have  minimum  voltage  across  the 
coupling  condenser.    Reversing  one  winding  reverses  the  sign  of  M . 


Mesh  6 


&  Combined  Self  and 
Mutual  Inductance 


F.  Inductance  and 
Capacitance  Coupling 


Mesh  GL 


Mesh6 


G.  Parallel  Resonant- 
Coupling 


H.  Mutual  Inductance  and 
Capacitance  Coupling 


FIG.  10.    Common  Types  of  Involved  Coupling 


5.  CURRENTS  AND  VOLTAGES  IN  COUPLED  CIRCUITS 


If  two  meshes  are  coupled  as  shown  ha  Fig.  11  the  primary  current  is 

«  B/z.' 


where 


and 


—  Taa  -f-  • 


(10) 
(lOo) 

(106) 


Note  that  in  these  equations  TM  and  r&&  are  the  self-resistances  of  the  respective  meshes 
(raa  =  TOO  +  rm}t  and  Xaa  and  XM  are  the  self -reactances.     (For  a  complete  discussion  of 


6-08 


PASSIVE   CIRCUIT  ELEMENTS 


self-impedances  and  mesh  currents  see  Section  5,  article  4.     zc'  is  called  the  equivalent 
primary  impedance  and  ra'  and  xd  the  equivalent  primary  resistance  and  reactance.    The 


Ll    '• 

1 

f     I»    } 

FIG.  11.    Two  Meshes  Coupled  through  an  Impedance  zm 

.second  term  on  the  right  of  eq.  (10a)  is  called  the  transferred  resistance,  and  the  correspond- 
ing term  of  eq.  (106)  the  transferred  reactance. 
The  secondary  current  is 

I&  =   Zmla/Zbb  =   ZmE/ZoaZ&'  (11) 

where 

/                    ,     Xm  Faa         Tm  ^aa         ^>fmXmXaa  /-,  -.     \ 

TV    =  ?bb  H o (Ha) 


and 


xbr 


—   XmXaa  ~ 


RESISTANCE  COUPLING.     When  the  coupling  is  a  pure  resistance,  eqs.  (lla  and  6) 
reduce  to 

Tb     =   rbb  -  rtr 

and 

Xb     =  ^66  +  T 

so  that  the  secondary  current  is 


and  its  absolute  value  is 
J6«  


r     ******  _L  Y   _L  r^^\  i 

Zoo     r&& =-  +  ;  I  Xbb  H S~  ) 

L  Zao2  \  2oa^    /J 


(13) 


w2roa\2 


(13o) 

;66  =  0)  is 
(14) 

REACTANCE  COUPLING.     When  the  coupling  is  a  pure  reactance,  eqs.  (lla  and  6) 
reduce  to 


The  maximum  value  possible  for  Ib  (for  optimum  tuning  arrangements,  xa 

T 

-i  &  max  max  : 


and 

Xb     =   Xbb  ~~  Xfn  Xaa/  Zaa, 

so  that  the  expression  for  the  secondary  current  becomes 
,  jxmE 


zoc 


and  its  absolute  value  is 


xm*rai 
zaa2 


./  Xm2Xaa\l 

3(Xbb-^^~)\ 


(156) 


(16) 


(16a) 


Since  the  nature  of  the  curve  of  secondary  current  against  mesh  reactances  (Fig.  12) 
changes  completely  at  the  point  where  xmz  —  raafbb  this  condition  is  defined  as  the  con- 
dition of  critical  coupling.  The  condition  when  xm2  >  raarbb  is  spoken  of  as  coupling 
greater  than  critical,  whereas  when  xmz  <  raor&&  the  coupling  is  said  to  be  less  than  critical. 


CURRENTS  AND  VOLTAGES  IN  COUPLED  CIRCUITS      6-09 


0  oca  -H 

FIG.  12.    Variation  of  J&  max  with  Primary  Reactance  for  Three  Degrees  of  Coupling 

These  conditions  have  been  variously  named  adequate  and  inadequate  coupling,  sufficient 
and  deficient  coupling,  super  and  sub  coupling;  there  is  no  general  agreement  as  to  termi- 
nology. 

When  the  coupling  is  less  than  or  equal  to  critical  there  is  only  one  inflection  point  on  the 
curve  of  /&  max  against  primary  reactance,  which  gives  a  maximum  value  for  xa  —  #&  =  0, 
or  when  each  mesh  is  separately  tuned  to  resonance.  When  the  coupling  is  greater  than 
critical  the  current  curve  shows  a  Tnim'Trm™  for  the  condition  xa  =  x&  =  0  with  maximum 
points  on  either  side  of  zero.  The  proper  adjustments  and  absolute  value  of  currents 
obtainable  are  shown  in  Table  1. 


Table  1. 


Conditions  for  and  Values  of  Maximum  Secondaty  Current  in  Two 
Mesh  Circuits  (For  Best  Possible  Tuning) 


Reactance  Coupling 

Resistance 
Coupling 

zm2  <raan& 

Xm*  =  TaaTbb 

Xmz  >Taarvb 

Opti'mTl'm  Yftlm^  for  Xa   -     -  *  • 

0 

0 

0 

•\J~(xm2-raar6&) 

Optimum  value  for  Xh       -  -  - 

0 

0 

0 

\l™  (xm*  -  Taar&ft) 
*Taa 

Maximum  secondary  current 

Exm 

E 

E 

Erm 

raarbb  4-  £m2 

2Vraarbb 

2Vr£MrZ)6 

Toanjb  -  Trr? 

Corresponding  primary  cur- 
cent         .    .          ........ 

Erw 

E 

E 

Eru> 

TooTM  +  Xm* 

2r<w 

2TcKj 

roan*  ~  rm* 

Coupled  circuits  are  frequently  used  to  match  two  circuits  with  dissimilar  impedances. 
The  conditions  listed  in  Table  1  for  maximum  secondary  current  also  give  the  conditions 
for  maximum  power  transfer  and  so  for  conjugate  impedances. 

TRANSMISSION-FREQUENCY  CHARACTERISTIC.  The  most-used  form  of  cou- 
pled circuit  is  that  of  Fig.  10,  case  H.  Assuming  that  the  distributed  capacitance  between 
windings  is  negligible,  the  transmission  formula  is 


H-  IP  -  (1  - 


(17) 


where  K  =  M/^/LiLz  is  called  the  coefficient  of  coupling,  and  di  and  d*  are  the  decrement 
coefficients  at  resonance  of  the  two  meshes.  Near  resonance  where  F  =  1  approximately 


r-  v== 


/r)//r]2 


(18) 


The  shape  of  the  transmission  curve  as  the  frequency  varies  depends  on  the  coefficients 
(di*  -f-  d£  —  2K?)  and  (didz  +  K^.  With  three  independent  variables  resulting  in  two 
coefficients  there  are  many  solutions  for  a  given  shape.  For  a  maximum  transmission 
the  additional  condition  may  be  specified  that  K  is  as  large  as  possible. 


6-10 


PASSIVE   CIRCUIT  ELEMENTS 


SELECTIVITY.  Comparing  eq.  (18)  with  eq.  (4)  for  a  single  circuit  it  is  seen  that  for 
frequencies  some  distance  from  resonance  the  transmission  for  a  coupled  circuit  varies 
roughly  inversely  as  the  square  of  the  departure  from  resonance,  while  for  the  single 
circuit  the  variation  is  roughly  as  the  inverse  first  power.  The  transmission  of  two  single 
circuits  in  cascade  (and  separated  by  a  vacuum  tube)  is  the  product  of  the  separate  trans- 
missions. For  two  circuits  of  decrement  coefficients  di  and  dz  this  would  be 


V[2</  - 


-  /r)//rP 


(19) 


Comparison  of  this  with  eq.  (18)  shows  that  the  selectivity  of  the  coupled  circuit  approaches 
that  of  the  cascaded  single  circuits  as  K  decreases,  but  the  transmission  of  the  coupled 

circuit  is  decreased  in  the  process. 
(For  complete  discussion  see  Puring- 
ton,  Proc.  I.R.E.,  Vol.  18,  983  [June 
1930],  and  Aiken,  Proc.  LR.E.,  Vol. 
25,  230  [February  1937].)  Figure  13 
shows  the  selectivity  to  be  expected 
with  well-designed  coupled  circuits. 
Curve  C  is  for  critical  coupling; 
curve  D  for  greater  than  critical 
coupling;  and  curves  A  and  B  for 
less  than  critical  coupling. 

STAGGERED  TUNING.  The 
selectivity  curve  of  a  pair  of  coupled 
resonant  circuits  may  be  duplicated 
by  the  use  of  a  two-stage  amplifier 
having  a  single  tuned  circuit  per 
stage,  one  circuit  being  tuned  above 
and  the  other  below  the  desired 
mean  frequency.  Similarly  any  de- 
sired number  of  single-tuned-circuit 
stages  may  be  used  to  obtain  an 
overall  selectivity  curve  with  a  flat 
top  and  steep  sides.  The  tuning 
points  are  distributed  systematically 
across  the  pass  bands.  The  effective 
Q  is  highest  on  the  circuits  tuned 
near  the  cutoff  frequency.  See  Sec- 
tion 7,  article  13;  also  the  literature 


_        _2Q  ;-  —  io  Q  10          20 

Frequency  Removed  from  Resonance  in  Kc.  Amplifiers 

FIG.  13.    Selectivity  of  Coupled  Circuits  «7,  critical  cou-  RCA  Rev.,  Vol.  V,  No.  3-4  [January- 

pling;  D,  greater  than  critical;  A  and  B,  less  than  critical)  April  1941])  . 


6.  AIR-CORE  TRANSFORMERS 

Since  almost  all  air-core  transformers  used  in  communication  circuits  employ  tuned 
secondaries  only  such  cases  will  be  considered.  A  simple  circuit  for  a  tuned  amplifier 
stage  is  shown  in  Fig.  14 A.  The  amplification  or  gain  of  such  a  stage  is  defined  as  the 
ratio  of  the  voltage  applied  to  the  grid  of  the  first  tube  to  the  voltage  delivered  to  the  grid 
of  the  second  tube.  The  value  of  the  gain  at  resonance  is 


Ei 


M 


(20) 


where  n  is  the  amplification  factor  of  the  tube. 

In  this  equation  (jPM*/rs  represents  the  impedance  reflected  into  the  primary  circuit 
by  the  tuned  secondary  circuit.  It  is  assumed  in  this  equation  that  the  primary  reactance 
is  negligible. 

This  equation  may  be  written  in  a  simpler  algebraic  form  but  is  most  easily  remembered 
and  used  in  the  form  shown.  Keeping  in  mind  that  (j^M^/rs  is  the  load  on  the  tube  it  is 

C^M^/TS 

evident  that  uEi  .  n^^ ,   N — ; is  the  voltage  drop  across  the  plate  load.    Multiplying 

(u*M*/ra}  +  rp 

this  voltage  by  the  ratio  of  transformation  L9/M  gives  JEfe,  the  output  voltage. 


AIR-CORE  TRANSFORMERS 


6-11 


As  the  mutual  reactance  between  the  primary  and  secondary  circuits  is  varied,  a 
maximum  of  gain  is  obtained  when  the  value  of  vf&P/rs  is  equal  to  rP.  This  constitutes 
matching  the  impedance  of  the  tube. 

In  modern  tubes  of  the  screen-grid  or  of  the  pentode  type,  the  plate  impedance  (rp)  is 
of  such  a  high  magnitude  that  it  is  ordinarily  impracticable  to  match  its  impedance.  In 
fact,  ordinarily  the  load  in  the  plate  circuit  is  so  small  as  to  have  a  negligible  effect  upon 

the  plate  current,  in  which  case  the  formula"  for  gain  becomes  ~  =    m          ~  ,  where 

EI  rs       M 

sm  is  the  mutual  conductance,  or  transconductance,  of  the  tube.  Expressing  this  equation 
in  words,  the  gain  of  a  tuned  amplifier,  employing  a  tube  of  very  high  plate  impedance, 
is  equal  to  the  transconductance,  multiplied  by  the  load  impedance,  and  by  the  trans- 
formation ratio. 


The  equation  may  be  written  in  another  form,  —  =  s, 


t  —  wJK". 


That  is,  the  gain 


is  equal  to  the  transconductance,  divided  by  the  power  factor  of  the  secondary,  and 
multiplied  by  the  mutual  reactance. 


FIG.  14.    Tuned  Amplifier  Circuits 


VARIATION  OF  GAIN  WITH  FREQUENCY  OF  RESONANCE.  The  gain  of  an 
amplifier  of  this  type  varies  considerably  with  frequency  as  it  is  tuned  over  the  frequency 
band.  The  frequency  squared  occurs  in  the  numerator,  causing  the  gain  to  tend  to  be 
larger  at  the  high-frequency  end  of  the  tuning  range.  This  tendency  is  partly  counter- 
balanced by  the  normal  variation  of  the  circuit  resistance  with  frequency.  The  rate  of 
change  of  resistance  varies  considerably  from  coil  to  coil,  but  the  resistance  varies  faster 
than  the  first  power  of  the  frequency,  and  slower  than  the  second  power.  If  the  resistance 
varies  directly  with  the  frequency  the  gain  is  proportional  to  the  frequency  of  resonance. 
However,  if  the  resistance  is  proportional  to  the  square  of  the  frequency  the  gain  is  con- 
stant. Usually  the  resistance  rises  only  slightly  faster  than  the  frequency,  causing  the 
gam  to  rise  with  frequency. 

EFFECT  OF  LEAKAGE  REACTANCE.  With  tight  coupling  it  is  justifiable  to  neglect 
the  primary  reactance  and  the  tube's  plate-filament  capacitance,  because  the  value  of  the 
reflected  impedance  exceeds  the  primary  reactance  so  greatly. 

However,  when  the  coupling  is  only  moderate,  the  plate-filament  capacitance  and  the 
primary  reactance  are  no  longer  negligible.  The  effect  is  to  increase  the  gain,  above  that 
of  the  formula,  especially  at  high  frequencies. 

In  practice,  the  transformer  is  frequently  of  such  a  design  that  the  primary  reactance 
is  far  from  negligible.  The  gain  formula  then  becomes  somewhat  too  complicated  for 
practical  use.  Cut-and-try  methods  are  the  rule  for  designing  such  transformers. 

TUNED  R-F  TRANSFORMER  EMPLOYING  COMPOUND  COUPLING.  A  com- 
mon design  of  transformer  in  broadcast  receivers  employing  screen-grid  tubes  is  that  shown 
in  Fig.  14B.  It  employs  a  primary  of  such  a  high  inductance  that  its  primary  is  resonant 
in  conjunction  with  the  output  capacitance  of  the  tube,  at  a  frequency  below  the  broadcast 
frequency  spectrum,  450  kc  and  below.  The  secondary  coil  is  of  the  usual  type,  but  the 
per  cent  coupling  is  quite  low,  usually  between  15  and  25  per  cent.  Sufficient  capacitance 
•coupling  is  used  to  give  the  desired  high-frequency  gain. 

The  performance  of  this  transformer  is  an  improvement  over  that  of  the  low-inductance 
primary  type,  in  that  the  gain  may  be  made  more  nearly  constant  throughout  the  tuning 
range. 

The  average  voltage  gain  in  this  type  of  transformer  is  somewhat  less,  but  the  great 
amplifying  ability  of  screen-grid  tubes  makes  it  unnecessary  to  obtain  the  greatest  possible 
gain  in  transformers. 


6-12 


PASSIVE  CIRCUIT  ELEMENTS 


7.  THREE-WINDING  TRANSFORMERS  (HYBRID  COILS) 

It  is  frequently  desirable  in  electric  circuits  that  currents  in  one  portion  of  a  circuit 
shall  induce  voltage  in  all  branches  of  the  circuit  except  certain  designated  ones,  in  which 
no  voltage  is  to  be  introduced.  This  may  be  accomplished  by  means  of  an  impedance 
bridge  (Fig.  15)  which  consists  of  six  adjustable  impedances  arranged  in  the  form  of  a 
Wheatstone  bridge.  If  the  impedances  are  so  adjusted  that  for  a  certain  frequency 
ZA/ZB  =  ZC/ZD  a  voltage  of  that  frequency  introduced  at  E  will  cause  no  current  in  the 
arm  Or,  and  vice  versa. 


FIG.  15.    Impedance  Bridge  Circuit 


FIG.  16.     Schematic  Circuit  of  Three- winding 
Transformer  with  Load 


A  more  widely  used  means  of  blocking  voltages  out  of  a  particular  branch  is  a  combina- 
tion of  three  coils  such  as  that  shown  in  Fig.  16;  it  is  optionally  called  a  three-winding 
transformer  or  a  hybrid  coil.  If  the  two  coils  La  are  wound  series  aiding  and  the  transformer 
is  well  made  so  that  the  winding  resistances  may  be  neglected  and  the  coefficients  of 
coupling  are  practically  unity,  then  a  voltage  E\  will  cause  no  current  through  ^2  (and 
vice  versa)  provided  z3  =  zJN1*,  where  TV  is  the  turn  ratio  between  one  of  the  La  coils  and 
the  Lb  coil.  Also  if  zi  =  22  a  voltage  #3  will  cause  no  current  through  E±,  and  vice  versa. 
If  all  these  conditions  are  fulfilled,  then  power  from  E\  or  E<z  will  divide  equally  between 
73  and  z4  and  power  from  E$  and  E±  will  divide  equally  between  zi  and  zj.  Such  trans- 
formers are  extensively  used  in  bidirectional  amplifiers  (see  p.  7-13) ;  also  this  circuit  is  the 
basic  circuit  for  all  neutralizing  circuits.  (See  p.  7-29) 


BIBLIOGRAPHY 

Adams,  J.  J.,  Undercoupling  in  Tuned  Coupled  Circuits  to  Realize  Optimum  Gain  and  Selectivity.. 

Proc.  I.R.E.,  Vol.  29,  277  (May  1941). 

Aiken,  C.  B.,  Two-mesh  Tuned  Coupled  Circuit  Filters,  Proc.  I.R.E.,  Vol.  25,  230  (February  1937). 
Blanchard,  J.,  The  History  of  Electrical  Resonance,  B.S.T.J.,  Vol.  20,  415  (October  1941). 
Everitt,  W.  LM  Communication  Engineering.    McGraw-Hill,  New  York  (1937). 

Johnson,  K  S.f  Transmission  Circuits  for  Telephonic  Communication.    Van  Nostrand,  New  York  (1925). 
King,  R.,  The  Application  of  Low-frequency  Circuit  Analysis  to  the  Problem  of  Distributed  Coupling 

in  Ultra-high-frequency  Circuits,  Proc.  I.R.E.,  Vol.  27,  715  (November  1939). 
King,  R.,  A  Generalized  Coupling  Theorem  for  Ultra-high-frequency  Circuits,  Proc.  I.R.E.,  Vol.  28,. 

84  (February  1940). 

Korman,  N.  I.,  Coupled  Resonant  Circuits  for  Transmitters,  Proc.  I.R.E.,  Vol.  31,  28  (January  1943). 
Landon,  V.  D.,  Cascade  Amplifiers  with  Maximal  Flatness,  R.C.A.  Review,  Vol.  5,  No.  3-4  (January 

1941). 

Lee,  R,,  A  Practical  Analysis  of  Parallel  Resonance,  Proc.  I.R.E.,  Vol.  21,  271  (February  1933). 
McElroy,  P.  K.,  Designing  Resistive  Attenuating  Networks,  Proc.  I.R.E.,  Vol.  23,  213  (March  1935). 
Mcllwain,  K.,  and  J.  G.  Brainerd,  High-frequency  Alternating  Currents.    John  Wiley,  New  York  (1939). 
Mead,  S.  P.,  Phase  Distortion  and  Phase  Distortion  Correction,  B.S.T.J.,  Vol.  7,  195  (April  1928). 
Nelson,  J.  R.,  A  Theoretical  Comparison  of  Coupled  Amplifiers  with  Staggered  Circuits,  Proc.  I.R.E.? 

Vol.  20,  1203  (July  1932). 

Purington,  E.  S.,  Single-  and  Coupled-circuit  Systems,  Proc.  I.R.E.,  Vol.  18,  983  (June  1930). 
Sherman,  J.  B.,  Some  Aspects  of  Coupled  and  Resonant  Circuits,  Proc.  I.R.E.,  Vol.  30,  505  (November 

1942). 

Terman,  F.  E.,  Radio  Engineering.    McGraw-Hill,  New  York  (1937). 
Zobel,  0.  J.,  Distortion  Correction  in  Electrical  Circuits,  B.S.T.J.,  Vol.  7,  438  (July  1928). 

Also  many  articles  in  Proc.  I.R.E.,  J.  A.I.E.E.,  and  Experimental  Wireless  and  Wireless  Eng. 


AUDIO-FREQUENCY  TRANSFORMERS  6-13 

TRANSFORMERS  WITH  IRON  CORES 

By  A.  J.  Rohner 

8.  AUDIO-FREQUENCY  TRANSFORMERS 

The  Function  of  the  audio-frequency  transformer  is  to  couple  various  circuits,  at  audio 
frequencies,  over  a  considerable  range  of  the  audio-frequency  band.  It  may  be  used  as 
an  impedance-matching  device,  as  a  means  of  isolating  circuits,  or  as  a  means  of  obtaining 
phase  reversal.  "When  it  couples  a  voltage  source,  such  as  a  microphone,  a  phonograph 
pick-up,  or  a  telephone  line,  to  the  grid  of  a  vacuum  tube,  it  is  usually  called  an  input 
transformer.  If  it  couples  the  plate  of  one  tube  to  the  grid  of  another,  it  is  called  an 
interstage  transformer.  If  it  couples  the  plate  of  a  vacuum  tube  to  some  sort  of  load, 
such  as  a  loudspeaker,  indicating  meter,  or  telephone  line,  it  is  referred  to  as  an  output 
transformer.  The  modulation  transformer  is  a  special  case  of  the  output  transformer,  in 
which  the  load  is  the  plate  of  a  radio-frequency  amplifier  tube.  If  the  transformer  is  used 
to  match  telephone  lines  of  unequal  impedance,  or  to  isolate  lines  of  equal  impedance, 
it  is  called  a  line  transformer.  There  are,  of  course,  many  variations  of  the  above-men- 
tioned applications. 

The  audio-frequency  transformer  is  constructed  much  like  a  power  transformer,  but 
there  are  two  distinct  points  of  difference.  Power  transformers  are  usually  one-frequency 
devices,  whereas  audio  transformers  must  operate  over  a  wide  band  of  frequencies.  In 
high-fidelity  systems,  for  example,  a  range  of  from  30  to  15,000  cycles  may  be  required. 
In  systems  where  intelligibility  of  speech  is  all  that  is  necessary,  300  to  3000  cycles  may 
suffice.  Then,  too,  the  power  transformer  works  from  a  voltage  source  having  good  regula- 
tion; that  is,  the  impedance  of  the  source,  or  "generator  impedance,"  is  negligible  as  com- 
pared with  the  load  impedance.  An  audio  transformer  always  works  from  a  voltage  source 
having  poor  regulation,  the  generator  impedance  often  being  equal  to  the  load  impedance, 
and  in  pentode  or  Class  B  power  amplifiers,  being  greater  than  the  load  impedance. 

These  two  factors,  wide  frequency  range,  and  high  generator  impedance,  place  severe 
restrictions  on  the  constants  of  an  audio  transformer.  Primary  inductance  must  be  high; 
leakage  inductance  and  distributed  and  other  ca- 
pacitances must  be  low.  See  G.  Koehler,  Design  of 
Transformers  for  Audio-frequency  Amplifiers  with 
Preassigned  Characteristics,  Proc*  LR.E.  (December 
1928);  P.  W.  Willans,  Low-frequency  Intervalve 
Transformers,  LE.E.  J.  (October  1926). 

COUPLED  CIRCUITS.  The  design  of  an  audio- 
frequency transformer  requires  the  solution  of 
coupled  circuits,  of  which  the  transformer  is  a  part, 
and  including  the  generator  and  the  load  imped-  FICL  i.  Fundamental  Circuit 

ances.    This  solution  may  be  carried  out  by  the 

classical  method,  using  mutual  inductance.    See  article  4,  Coupled  Circuits.    Thus,  for 
the  circuit  of  Fig.  1,  the  voltage  impressed  on  the  primary  is 


CD 
TL  -r  J^^a 

The  impedance  "looking  in"  to  the  primary  is 

.SH/rt, 

(2) 


The  voltage  across  the  load,  TL,  is 

Et  =  rzJ2  =     TV*,  (3) 


This  method  becomes  quite  complicated,  however,  when  all  the  factors  affecting  audio 
transformers  are  considered:  core  loss,  winding  resistances,  distributed  capacitance,  inter- 
winding  capacitance,  turns  ratio,  etc.  The  first  step  in  simplifying  the  analysis  of  the 
audio  transformer  and  its  associated  circuits  is  to  convert  them  to  an  equivalent  direct- 
connected  network.  See  J.  H.  Moreeroft,  Principles  of  Radio  Communication,  2nd  Ed., 
pp.  95-105. 

EQUIVALENT  DIRECT-CONNECTED  NETWORK.  Considering  the  transformer 
shown  in  Fig.  1  as  a  unity-ratio  transformer,  LI  —  £2.  Then  kLi  =  kL$,  where  k  » 
coefficient  of  coupling,  and  M  =  fcVZiLa  —  kLi,  where  M  is  the  mutual  inductance. 


6-14 


PASSIVE   CIRCUIT  ELEMENTS 


The  leakage  inductance  of  the  primary,  or  that  portion  of  the  primary  which  is  not 
coupled  to  the  secondary,  equals  (1  —  &)Z/i,  Similarly,  the  portion  of  the  secondary  not 
coupled  to  the  primary  equals  (1  —  fc)La»  which  is,  of  course,  equal  to  (1  —  k)Li.  The 
circuit  of  Fig.  1  can  now  be  replaced  by  a  direct-connected  network  as  shown  by  Fig.  2. 
The  common  impedance  is  kLi. 

It  may  be  shown  that  this  circuit  is  the  exact  equivalent  of  Fig.  1.  For  example,  the 
input  impedance  equals  Z\  and 


Substituting  M  for  kLi,  and  1/2  for  LI 


TL  4-  jwLi 
to'Jif8 

?"L  +  juLiz 


which  is  the  expression  derived  for  the  circuit  of  Fig.  1. 

The  equivalent  network  of  an  inequality  ratio  transformer  may  be  referred  to  either 
the  primary  or  secondary  circuit,  provided  the  correct  transformations  are  made.    To  reflect 
.  n  the  secondary  constants  to  the  primary, 

P        u-AOLi.     U-*;L2  for  example?  it  is  necessary  to  multiply 

all  the  impedances  on  the  secondary  side 
by  the  ratio  Li/Z/2-  With  iron-core  trans- 
formers, the  ratio  1/1/1/2  is  for  all  practical 
purposes  equal  to  the  square  of  the  turns 
ratio.  Voltages  are  transformed  by  the 
turns  ratio,  currents  by  the  inverse  of  the 
turns  ratio, 
secondary  turns,  the  secondary  constants,  referred 


FIG.  2.    Equivalent  Network 

If  Np  =  primary  turns,  and  N8  = 
to  the  primary,  are 


After  these  conversions  are  made,  the  transformer  becomes  a  unity-ratio  transformer 
and  can.  be  replaced  by  an  equivalent  direct-connected  network. 

COMPLETE  EQUIVALENT  NETWORK.  Figure  3  shows  the  complete  equivalent 
network  of  an  audio-frequency  transformer,  referred  to  the  primary,  in  which  EI  is  the 
generator  voltage  (pEg  in  the  case  of  a  vacuum  tube);  rp  is  the  generator  resistance; 
Ci  is  the  distributed  capacitance  of  the  primary  winding,  plus  any  additional  capacitance 
across  that  winding;  TI  is  the  primary  winding  resistance;  rc  is  the  core-loss  resistance;  r-^f 
is  the  secondary  winding  resistance, 

referred  to  the  primary;  C2'  is  the  .  TP         .ri    <l-fc>Li 

distributed  capacitance  of  the  sec- 
ondary winding,  plus  additional 
capacitances  across  that  winding 
within  the  transformer  itself,  re- 
ferred to  the  primary;  and  rz,',  Ci/, 
and  JSV  are  the  load  resistance,  load 
capacitance,  and  load  voltage,  all 
referred  to  the  primary. 


FIG.  3.    Equivalent  Network  of  Audio  Transformer 


The  "additional  capacitances"  spoken  of  may  be  capacitances  of  the  windings  to  ground, 
or  capacitance  between  windings.  For  example,  a  single-ended  interstage  transformer 
might  have  its  primary  on  the  inside,  next  to  the  core,  and  secondary  on  the  outside, 
wound  over  the  primary.  If  the  primary  finish  is  connected  to  +B,  and  the  secondary 
start  is  connected  to  ground,  there  is  no  a-c  potential  between  the  adj  acent  surfaces  of  the 
two  windings,  so  that  capacitance  between  the  two  windings  has  no  effect.  However,  the 
primary  start  would  go  to  plate,  and  the  start  layer  of  the  winding  has  capacitance  to  the 
core.  This  capacitance  is  an  additional  capacitance  across  the  primary  winding.  Simi- 
larly, the  secondary  finish  would  go  to  grid,  and  any  capacitance  existing  between  the 
finish  layer  of  the  secondary  and  the  core  or  case  would  be  an  additional  capacitance  across 
the  secondary  winding.  If  the  connections  to  both  windings  were  reversed,  making  primary 
start  +J5,  and  secondary  finish  ground,  those  capacitances  would  have  no  effect.  The 
capacitance  between  windings,  however,  would  be  of  great  importance.  This  would  be 
somewhat  equivalent  to  an  additional  capacitance  across  the  winding  having  the  most 
turns.  Use  of  a  grounded  electrostatic  shield  between  windings  will  largely  eliminate 


AUDIO-FREQUENCY  TRANSFORMERS 


6-15 


capacitance    between    windings,    but,    of    course,    it    adds    additional    capacitances    to 
ground. 

Solution  of  the  circuit  of  Fig.  3  will  give  the  characteristics  of  the  audio  transformer, 
such  as  (1)  voltage  ratio  as  a  function  of  frequency;  (2)  primary  impedance  as  a  function 
of  frequency;  (3)  phase  shift  as  a  function  of  frequency;  (4)  efficiency. 

SIMPLIFIED  NETWORK  AT  LOW  FREQUENCIES.  The  network  of  Fig.  3  can  be 
greatly  simplified  for  practical  design  purposes  by  considering,  first,  the  effect  of  frequencj^ 
upon  the  relative  importance  of  the  various  constants,  and  second,  the  particular  applica- 
tion in  which  the  transformer  is  used. 

At  low  frequencies,  leakage  inductance 
and  all  the  capacitances  can  be  ignored. 
Furthermore  the  coefficient  of  coupling,  k, 
of  most  audio  transformers  is  above  0.995, 
so  that  kLi  may  be  taken  as  LI  with  an 
error  of  less  than  1  per  cent.  The  low- 
frequency  characteristics  of  any  audio 
transformer  are  determined  by  the  pri- 
mary inductance  and  the  various  resist- 
ances of  the  network,  as  shown  in  Fig.  4. 

But  Fig.  4  can  be  simplified  still  further, 
single  resistor,  as  shown  in  Figs.  5  and  6. 


FIG.  4.    Network  at  Low  Frequencies 


The  resistances  can  be  lumped  together  into  a 
Let  ra  =  rp  +  r\  and  let  r&  =  rj  -f  ^i/-    Then 
the  lumped  resistance,  R,  is  equivalent  to  ra,  r&,  and  re  in  parallel,  or 


R  = 


(4) 


If  the  input  voltage,  EI,  is  multiplied  by  the  attenuation  of  the  resistance  network,  the 
circuit  of  Fig.  5  becomes  identical,  in  its  output  voltage  and  phase  shift,  with  Fig.  4. 
The  new  value  for  the  input  voltage,  of  Fig.  5,  is 


E  = 


(5) 


4-  Tofc  +  TbTc 

Figure  6  is  simply  a  redrawing  of  Fig.  5,  showing  the  single  resistor  and  the  new  input 
voltage.    Figure  6  is  an  exact  equivalent  of  Fig.  4  as  far  as  voltage  output  and  phase 


Eix  rarb+rarc+rbrc 


FIG.  5.     Method  of  Combining  Resistances 


FIG.  6.    Circuit  of  Audio  Trans- 
former at  Low  Frequencies 


shift  are  concerned.  The  voltage  output  and  the  phase  shift,  at  low  frequencies,  of  all 
audio  transformers  are  given  by  this  simple  circuit.  Figure  7  shows  those  characteristics, 
as  a  function  of  27rfLi/R. 

At  some  frequency,  2irfLi  =  Rt  so  that  2irfLi/R  =  1-  Let  this  particular  frequency  be 
called  /i.  This  is  the  low  frequency  at  which  the  response  has  dropped  3  db  from  its  value 
in  the  middle-frequency  range.  At  twice  this  frequency,  2ir/Li/R  =  2.  At  three  times 
this  frequency,  2irfLi/R  =  3,  etc.  At  any  frequency,  /,  2irfLi/R  =  f/J\.  The  curves 
given  in  Fig.  V,  as  a  function  of  2irfLi/R,  are  at  the  same  time  frequency  characteristics, 
the  frequency  being  expressed  in  terms  of  the  reference  frequency,  f\. 

In  order  to  determine  the  proper  value  of  primary  inductance,  it  is  necessary  to  know 
what  drop  in  secondary  voltage  is  permissible,  at  some  specified  low  frequency,  as  com- 
pared with  the  voltage  in  the  middle-frequency  range.  For  example,  1-db  drop  at  100 
cycles  might  be  given  as  the  requirement  of  low-frequency  response.  From  Fig.  7,  the 
ratio  of  2irfLi/R  is  found,  which  gives  this  particular  drop.  As  a  first  approximation,  the 
winding  and  core-loss  resistances  may  be  neglected,  so 

(6) 
(7) 


"  —  Tv  +  rL' 


Then 


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Ratio  Primary  Reactance/Resistance=2^/lD1/K.  or  Frequency  Ratio—///! 
FIG.  7.    Response  and  Phase  Shift  at  Low  Frequencies 

SIMPLIFIED  NETWORK  AT  MIDDLE  FREQUENCIES.  In  this  frequency  range, 
all  reactance  elements  become  negligible,  and  the  transformer  reduces  to  a  network  of 
resistances,  as  shown  in  Fig.  8.  In  this  range,  phase  shift  is  practically  zero.  This  is  the 
"flat"  portion  of  the  frequency-response  characteristic,  the  secondary  voltage  being 


-  N*  v 

77~  X 


v 

X 


TL'Te 


; 
-f-  rarc 


and  the  efficiency  of  the  transformer  being,  to  a  very  close  approximation, 


Efficiency  = 


rL' 


2r2' 


(8) 


(9) 


SIMPLIFIED  NETWORK  AT  HIGH  FREQUENCIES.     The  shunting  effect  of  the 

primary  inductance  is  negligible  at  high  frequencies.  See  Fig.  7.  If  the  generator  resist- 
ance, rp,  or  the  reflected  load  resistance,  r^/,  is 
less  than  20,000  ohms,  the  primary  capaci- 
tance, Ci,  may  be  neglected.  Most  audio 
transformers  fall  in  this  class.  Similarly,  con- 
sidering the  secondary  side  of  the  transformer, 
if  the  reflected  generator  resistance,  or  the 
load  resistance,  is  less  than  20,000  ohms,  the 
secondary  capacitance,  Ca,  may  be  neglected. 

FIG.  8.     Equivalent  Network  at  Middle  Fre-    Though  this  is  usually  true  of  output  trans- 
quencies  formers,  it  is  seldom  true  of  input  or  interstage 

transformers. 
The  core-loss  resistance,  rc,  has  little  effect  at  high  frequencies  beyond  reducing  the 

secondary  voltage  by  a  few  per  cent.    The  per  cent  voltage  drop  caused  by  core  loss,  using 

the  symbols  of  Fig.  8,  is 


Core  loss  drop 


100 


rc/ra 


per  cent 


(10) 


This  usually  amounts  to  2  or  3  per  cent.    As  far  as  the  shape  of  the  response  curve,  or 

the  amount  of  phase  shift,  is  concerned, 

core  loss  may  be  neglected.  rp  rv      2(l-fc)L1       r2' 

Neglecting  primary  inductance,  pri-  ^^r^AA/V — A/W^ — \J&SLQSL> — MAA/- 
mary  capacitance,  and  core  loss,  the 
equivalent  circuit,  at  high  frequencies, 
becomes  as  shown  in  Fig.  9.  The  term 
2(1  —  K)Lit  is  called  the  leakage  induct- 
ance referred  to  the  primary  and  is  usu- 
ally designated  by  Le.  FIG.  9.  Equivalent  Network  at  High  Frequencies 


OUTPUT  TRANSFORMERS 


6-17 


9.  OUTPUT  TRANSFORMERS 

The  function  of  the  output  transformer  is  to  transfer  power  from  the  plate,  or  plates, 
of  vacuum  tubes  to  a  load,  such  as  a  loudspeaker,  an  indicating  meter,  or  a  line.  It  provides 
the  necessary  impedance  transformation,  and  it  isolates  the  load  from  the  d-c  potential 
and  current  of  the  plate  circuit.  Efficiency  is  usually  important,  and  the  transformer 
must  meet  a  prescribed  frequency-response  characteristic. 

TURNS  RATIO  is  determined  by  the  plate  load  recommended  for  the  tube,  or  tubes, 
by  the  tube  manufacturer,  rj/,  and  the  actual  load  resistance,  r&. 

(11) 

FREQUENCY  RESPONSE  is  controlled  by  the  amount  of  primary  inductance,  at  low 
frequencies,  and  by  the  amount  of  leakage  inductance,  at  high  frequencies.  The  allowable 
drop  at  low  frequencies  fixes  a  rninimuin  value  of  primary  inductance.  See  Fig.  7  and 
eqs.  (6)  and  (7). 

The  equivalent  network  of  the  output  transformer  at  high  frequencies  is  given  by  Fig. 
10,  which  is  the  same  as  Fig.  9  except  that  all  capacitances  have  been  omitted.  The 


i — vjQJL<t» — i ' — r™ 

T  <  >/ 

r        R^**^  f2 


FIG.  10.    Output  Transformer  at  High  Frequencies 


FIG.  11.     Circuit  Equivalent  to  Fig.  10 


capacitance  of  an  ordinary  audio  transformer  winding  will  be  about  100  ju^i/,  more  or  less, 
depending  upon  the  coil  construction.  At  10,000  cycles,  this  is  160,000  ohms  of  capacitive 
reactance.  The  load  on  an  output  transformer  is  seldom  over  a  few  thousand  ohms  and 
may  be  as  low  as  3  or  4  ohms.  The  shunting  effect  of  the  secondary  capacitance  is  there- 
fore negligible,  even  at  the  highest  audio  frequencies.  The  reflected  load,  looking  into 
the  primary,  is  seldom  greater  than  20,000  ohms,  e.g.,  pushpull  6F6  tubes  require  10,000 
ohms.  Again,  primary  capacitance  is  negligible. 

The  circuit  of  Fig.  11  is  the  exact  equivalent  of  that  of  Fig.  10,  as  far  as  voltage  output 
and  phase  shift  are  concerned.     The  voltage  output  and  the  phase  shift  of  all  output 


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Ratio  Leakage  Reactance /Resistance  —  ^fLe/Rf  or  Frequency  Ratio=//f2 
FIG.  12.    Response  and  Phase  Shift.    Output  Transformers  at  High  Frequencies 

transformers  at  high  frequencies  are  given  by  this  simple  circuit,  provided  that  the  load 
is  resistive  and  the  load  and  reflected  load  are  less  than  20,000  ohms.  Figure  12  shows 
those  characteristics  as  a  function  of  2irfLe/R. 


6-18  PASSIVE   CIRCUIT  ELEMENTS 

At  some  frequency,  27rfLe  —  R,  so  that  2TrfLe/R  =  1.  Call  this  particular  frequency /2. 
This  is  the  high  frequency  at  which  the  response  has  dropped  3  db  from  its  value  in  the 
middle-frequency  range.  At  twice  this  frequency  2irfLe/R  =  2.  At  three  times  this 
frequency,  2irfLe/R  =  3,  etc.  At  any  frequency,  /,  27rfLe/R  —  f/h.  The  curves  given 
in  Fig.  12,  as  a  function  of  2x/Le/Jf2,  are  at  the  same  time  frequency  characteristics,  the 
frequency  being  expressed  in  terms  of  the  reference  frequency,  /2. 

To  determine  the  allowable  amount  of  leakage  inductance,  it  is  necessary  to  know  what 
drop  in  secondary  voltage  is  permissible,  at  some  specified  high  frequency,  as  compared 
with  the  voltage  in  the  middle-frequency  range.  From  Fig.  12,  the  ratio  of  2irfLe/R 
which  gives  this  drop  is  found. 

R  =  rp  +  n  4-  JM'  +  rL' 
Then 

2irfLe          R 

^--B     XW  (12) 

See  F.  E.  Terman,  Radio  Engineering,  2nd  Ed.,  pp.  293-299;  L.  A.  Kelley,  Transformer 
Design,  Rad.  Engrg.  (December  1934,  February  1935) ;  F.  E.  Terman  and  R.  E.  Ingebret- 
sen,  Output  Transformer  Response,  Electronics,  January  1936;  and  Magnetic  Circuits  and 
Transformers,  staff  of  M.I.T.,  pp.  472-486. 

EFFICIENCY  of  output  transformers  is  usually  between  80  and  90  per  cent,  though  it 
may  be  as  low  as  60  per  cent  for  cheap  or  poorly  designed  transformers.  Maximum 
efficiency  is  obtained,  for  a  given  physical  size  and  core  material,  when  copper  loss  =  core 
loss,  and  when  primary  copper  loss  =  secondary  copper  loss.  Such  a  balance  of  losses  is 
not  always  possible,  however.  If  the  secondary  resistance,  r%,  is  made  5  per  cent  of  the 
load  resistance,  rj,;  if  the  primary  resistance,  n,  is  made  5  per  cent  of  the  reflected  load 
resistance,  rif;  and  if  the  core-loss  resistance,  rc,  is  made  10  times  the  reflected  load,  ri/, 
the  losses  will  be  very  nearly  balanced,  and  the  efficiency  will  be  82  per  cent.  See  eq.  (9) . 

LOUDSPEAKER  LOAD.  The  analysis  of  output  transformers  given  above  has 
assumed  a  constant,  resistive  load.  If  the  load  impedance  is  not  constant  throughout  the 
frequency  range,  the  frequency-response  characteristic  will  not  be  flat  but  will  rise  and 
fall  where  the  load  impedance  rises  and  falls.  This  effect  is  not  very  pronounced  with 
Class  A  triode  amplifiers.  However,  with  pentode  or  with  Class  B  triode  amplifiers,  the 
output  voltage  is  approximately  proportional  to  the  load  impedance.  When  such  tubes 
are  used  to  drive  the  conventional  moving-coil  loudspeaker,  a  flat  frequency-response 
characteristic  is  no  longer  obtained. 

The  impedance-frequency  characteristic  of  a  moving-coil  loudspeaker  is  characterized 
by  a  low-frequency  resonance  peak  and  by  a  rise  in  impedance  with  frequency  at  the 
higher  audio  frequencies.  In  the  neighborhood  of  400  cycles,  the  loudspeaker  impedance 
is  the  minimum  and  is  resistive.  This  minimum,  resistive  impedance  should  be  used  as 
the  basis  of  calculating  the  turns  ratio  and  the  efficiency  of  the  transformer. 

Flattening  of  the  frequency-response  curve  by  mismatching,  that  is,  by  using  a  value 
for  rz,  perhaps  twice  the  value  of  the  actual  minimum,  resistive  impedance,  in  order  to 
favor  the  low  and  high  frequencies  at  the  expense  of  the  middle  frequencies,  is  somewhat 
effective  when  used  with  Class  A  triode  amplifiers.  It  does  not  level  the  voltage  char- 
acteristic but  does  tend  to  level  the  power  output  over  a  wider  frequency  range.  Mis- 
matching is  futile,  however,  when  used  with  pentode  or  Class  B  amplifiers. 

The  rise  in  impedance  at  the  high  frequencies  may  be  offset  by  connecting  a  capacitor, 
or  a  resistor  and  capacitor  in,  series,  across  the  primary  of  the  output  transformer.  The 
low-frequency  peak  is  best  controlled  by  acoustical  damping  of  the  loudspeaker  itself. 

PTTSHPULL  OUTPUT  TRANSFORMER,  CLASS  A.  No  special  problems  are  intro- 
duced by  pushpull  operation  of  the  output  transformer  if  the  amplifier  is  Class  A.  (See 
Section  7,  Amplifiers.)  The  generator  resistance,  rp,  is  twice  the  plate  resistance  of  one 
tube.  The  reflected  load,  rift  is  the  recommended  tube  load,  plate-to-plate.  In  fact,  the 
design  of  the  transformer  is  simpler,  for  pushpull  operation,  because  the  d-c  plate  currents 
of  the  two  tubes  flow  in  opposite  directions  in  the  transformer  windings,  so  that  d-c 
magnetization  of  the  core  is  canceled  out.  This  results  in  higher  primary  inductance  and 
a  better  low-frequency  response. 

Data  are  furnished  by  the  tube  manufacturers  on  optimum  plate-to-plate  load.  The 
method  of  arriving  at  this  optimum  load  is  discussed  by  B.  J.  Thompson,  Graphical  Deter- 
mination of  Performance  of  Pushpull  Audio  Amplifiers,  Proc.  I.R.E.,  April  1933. 

PUSHPULL  OUTPUT  TRANSFORMER,  CLASS  B.  Class  B  operation  imposes 
special  requirements  on  the  output  transformer,  because  one  half  of  the  primary  works 
during  one  half-cycle,  and  the  other  during  the  other  half-cycle.  It  is  important  that  the 
two  halves  of  the  primary  be  closely  coupled,  so  that  the  cross-over  from  one  to  the  other 
may  be  accomplished  smoothly  and  without  introducing  transients.  It  is  also  important 
that  each  half  of  the  primary  be  coupled  equally  to  the  secondary.  Otherwise  the  high- 


INPUT  AND  INTERSTAGE  TRANSFORMERS 


6-19 


FIG.  13.    Arrangement  of  Windings,  Clas 
B  Output  Transformer 


frequency  response  of  the  transformer  will  not  be  the  same  for  both  half-cycles,  which 
will  produce  even  harmonics  in  the  output  wave.  In  general,  leakage  inductance  should 
be  kept  to  the  minimum  between  the  windings  of 
a  Class  B  output  transformer,  even  beyond  the  re- 
quirements of  frequency  response. 

These  requirements  of  low  leakage  and  equal  coup- 
ling are  met  by  using  the  coil  construction  illustrated 
in  Fig.  13. 

THE  MODULATION  TRANSFORMER  is  an 
output  transformer,  which  has  as  its  load  the 
plate  of  a  Class  C  radio-frequency  amplifier.  This 
load  is  resistive  and  is  usually  of  the  order  of  a  few 
thousand  ohms.  (See  Section  7,  Modulators.)  The 
audio-frequency  generator  is  usually  a  Class  B  am- 
plifier, so  that  the  discussion  of  Class  B  output  transformers  given  above  applies  to  mod- 
ulation transformers. 

The  secondary  often  is  required  to  carry  the  d-e  current  of  the  Class  C  amplifier.  This 
produces  a  d-c  magnetization  of  the  core,  which  must  be  considered  when  designing  the 
transformer,  because  of  its  effect  upon  inductance  and  low-frequency  response  as  well  as 
upon  heating.  Core  saturation,  due  to  d-c  and  a-c  magnetization,  is  usually  an  important 
factor  in  modulation  transformers.  It  is  most  serious  at  the  lowest  frequency  of  the  fre- 
quency range,  since  there  the  a-c  flux  density  is  greatest.  The  effects  of  d-c  magnetization 
of  the  core  upon  inductance  and  upon  saturation  are  discussed  in  Section  3,  Ferrous-cored 
Inductors. 

Modulation  transformers  often  work  at  high  power  levels,  of  the  order  of  hundreds  or 
thousands  of  watts.  Heating  is  an  important  consideration,  as  with  power  transformers. 
Usually,  too,  high  voltages  are  applied  to  the  modulation  transformer.  The  primary 
center  tap  and  one  end  of  the  secondary  winding  are  connected  to  the  d-c  plate  supply 
and  must  be  insulated  to  withstand  its  voltage.  The  ends  of  the  primary  and  the  other 
end  of  the  secondary,  all  of  which  are  connected  to  plates,  must  withstand  twice  the  d-c 
plate  supply  voltage,  since  they  have  audio-frequency  voltage,  additional  to  the  d-c 
voltage,  and  of  a  peak  value  approximately  equal  to  the  d-c  voltage. 

Since  the  primary  center  tap  and  one  end  of  the  secondary  are  often  connected  to  a 
common  point,  the  B  supply,  and  since  the  load  resistance  of  the  Class  C  amplifier  is  of 
the  same  order  of  magnitude  as  the  reflected  load  on  one  tube  of  the  Class  B  amplifier,  there 
is  a  strong  temptation  to  make  the  modulation  transformer  an  auto  transformer,  having 
one  half  of  the  primary  common  with  the  secondary.  The  saving  in  size,  cost,  and  insula- 
tion by  such  construction  is  very  great.  Invariably,  however,  such  construction  leads  to 
trouble  due  to  the  unequal  coupling  between  the  two  halves  of  the  primary  and  the  sec- 
ondary. (See  above,  Pushpull,  Class  B.) 

For  a  practical  analysis  of  Class  B  modulation,  and  driver,  transformers,  see  J.  Kunz, 
Transformers  for  Class  B  Modulators,  Radio  Engrg.,  July  1934. 

THE  LINE  TRANSFORMER  is  not  an  output  transformer,  as  it  does  not  work  out  of 
the  plate  of  a  vacuum  tube.    However,  the  analysis  presented  above  for  the  output  trans- 
former applies  equally  well  to  the  line  transformer. 
Turns  ratio  is  determined  in  the  same  manner.    Fre- 
quency response  and  phase  shift  are  governed  by  the 
same  factors,  viz.,  primary  and  leakage  inductances. 
Line  transformers  usually  operate  at  low  power  lev- 
els, of  the  order  of  6  milliwatts,  or  less,  so  that  shield- 
ing from  stray  magnetic  fields  may  be  necessary. 

Often,  too,  it  is  necessary  to  balance  the  line  to 
ground.    This  means  that  the  capacitance  from  one 

FIG.  14.    Balanced  Coil  Construction     end  of  the  primary  to  ground  shall  equal  that  from  the 

other,  and  the  capacitance  from  one  end  of  the  second- 
ary to  ground  shall  equal  that  from  the  other.  A  symmetrical  coil  construction,  as  shown 
in  Fig.  14,  accomplishes  this  purpose. 


10.  INPUT  AND  INTERSTAGE  TRANSFORMERS 

The  function  of  the  input  transformer  is  to  couple  an  audio-frequency  voltage  source, 
such  as  a  microphone,  phonograph  pick-up,  or  telephone  line,  to  the  grid  of  a  vacuum  tube. 
That  of  the  interstage  transformer  is  to  couple  the  plate  of  one  tube  to  the  grid  of  another. 
Either  type  must  conform  to  a  predetermined  frequency-response  characteristic  and  must 
furnish  the  greatest  possible  voltage  amplification  consistent  therewith. 


6-20  PASSIVE   CIRCUIT  ELEMENTS 

With  either  type  of  transformer,  the  load  consists  of  the  grid  circuit  of  a  vacuum  tube 
(or  tubes),  and  its  impedance  is  very  high,  frequently  of  the  order  of  megohms.  It  is 
often  no  more  than  the  input  capacitance  of  the  tube,  although  sometimes  a  resistor  of 
100,000  to  500,000  ohms  is  placed  across  the  secondary,  also.  With  such  a  high-impedance 
load,  the  secondary  winding  capacitance  is  not  negligible.  The  secondary  winding  capaci- 
tance and  the  load  capacitance,  together,  largely  control  the  high-frequency  response  and 
the  turns  ratio  of  the  transformer. 

FREQUENCY  CHARACTERISTICS,  LOW  AND  MIDDLE  FREQUENCIES.  The 
same  analysis  applies  to  input  and  interstage  transformers,  at  these  frequencies,  as  applies 
to  other  audio  transformers.  See  above,  Simplified  Network  at  Low  Frequencies  and 
Middle  Frequencies.  If  there  is  no  resistance  load  on  the  secondary,  that  is,  if  rif  =  oo , 
the  equations  given  become  simpler.  Thus,  eq.  (8),  for  the  secondary  voltage,  becomes 


and  eq.  (9) ,  for  efficiency,  becomes 

Efficiency  =  0  (14) 

FREQUENCY  CHARACTERISTIC,  HIGH  FREQUENCIES.  Figure  9  shows  the 
equivalent  network  at  high  frequencies.  Considering  first  the  case  when  there  is  no 
secondary  resistance  load,  the  equivalent  circuit  becomes  that  shown  in  Fig.  15.  This  is 
a  simple  series  resonant  circuit.  It  is  convenient  to  express  the  performance  of  this  circuit 
by  means  of  a  family  of  curves,  plotting  voltage  ratio  vs.  frequency,  as  shown  by  Fig.  16. 
It  is  seen  that  the  shape  of  the  desired  frequency-response  characteristic  determines  the 

value  of  a  constant,  N,  while  the  position  of  the 
rp          r  U,  rr  desired  characteristic   on   the    frequency  band 

2  determines  the  value  of  the  resonant  frequency, 

/o.    Then 

£j  =  N(rp  +  n  +  raO  (15) 


FIG.  15.    Input  or  Interstage  Transformer  at        _  ,  .  ,,-,,, 

High  Frequencies  From  these  equations,  the  value  of  the  leakage 

inductance,  Le,  and  of  the  reflected  secondary 

capacitances,  C',  which  will  give  the  desired  frequency  response,  can  be  calculated.  If  the 
amplification  at  high  frequencies  is  to  be  substantially  constant,  the  value  of  the  cpnstant 
N  must  lie  between  0.75  and  1.0.  See  F.  E.  Terman,  Radio  Engineering,  2nd  Ed.,  pp. 
188-202;  Gen.  Elec.  Tech.  Report  19366,  Audio  Transformer  Design  (July  1930);  and 
Magnetic  Circuits  and  Transformers,  staff  of  M.I.T.,  pp.  486-494. 

Considering  the  case  in  which  a  resistance  load  is  placed  across  the  secondary,  in  addition 
to  the  tube  input  and  secondary  winding  capacitances,  the  equivalent  network  is  that  of 
Fig.  9.  The  resistance  load  will  reduce  the  leakage  resonance  peak,  at  the  same  time  lower- 
ing the  voltage  output  in  the  middle-frequency  range  in  accordance  with  eq.  (8) .  How- 
ever, the  effect  is  more  pronounced  in  the  region  of  leakage  resonance,  so  that  an  overall 
flattening  of  the  response  characteristic  results.  See  Terman,  p.  199. 

No  single  family  of  curves  can  be  drawn  which  will  show  the  performance  of  this  circuit. 
However,  two  methods  of  attacking  the  problem  have  been  described.  One  method  uses  a 
master  chart  showing  the  response  at  the  leakage-resonance  frequency.  This  chart 
enables  a  designer  to  pick  out  a  value  of  secondary  loading  resistance  that  will  give  the 
desired  response  at  resonance.  He  then  computes  the  response  at  a  few  other  points, 
sufficient  for  plotting  the  frequency-response  characteristic.  See  P.  W.  Klipsch,  A.F. 
Amplifier  Circuits  Using  Transformers,  Proc.  I.R.E.,  February  1936. 

Another  method  of  solving  the  circuit  of  Fig.  9  is  to  draw  several  families  of  curves, 
each  family  representing  some  fixed  relationship  between  the  various  constants.  This 
method,  with  six  such  families  of  curves,  is  described  by  J.  G.  Story,  Design  of  A.F.  Input 
and  Interval ve  Transformers,  Wireless  Engr.,  February  1938. 

THE  TURNS  RATIO,  of  an  input  or  interstage  transformer  is  determined  by  the 
secondary  capacitance,  that  is,  by  the  sum  of  the  secondary  winding  capacitance,  (?2,  and 
the  input  capacitance  of  the  tube,  CL-  Call  this  total  capacitance  C,  and  let  its  value 
reflected  to  the  primary  be  C'.  The  correct  value  of  Q'  is  found  from  the  frequency- 
response  requirements,  as  described  above.  Then 


INPUT  AND  INTERSTAGE  TRANSFORMERS 


6-21 


The  capacitance,  distributed,  and  to  ground,  of  the  secondary  winding  cannot  be 
calculated  accurately  until  the  design  of  the  transformer  has  been  completed.  As  a  first 
approximation,  for  finding  the  turns  ratio,  a  value  of  50  MM/  may  be  used.  The  input 
capacitance  of  the  tube  is  given  by  the  formula 

CL  «  Cgf  +  Cfp(l  4-  M) 

where  Cgf  —  static  capacitance  between  grid  and  filament,  Cgp  —  static  capacitance  be- 
tween grid  and  plate,  and  M=  effective  amplification  of  tube. 


.4      .5     .6        .8 

Frequency 
PIG.  16.    Leakage  Resonance 

As  an  example  of  what  can  be  expected  in  the  way  of  turns  ratio,  assume  that  all  sec- 
ondary capacitances  total  100  MMf  and  that  the  high-frequency  response  of  the  trans- 
former is  as  given  by  N  =  1  of  Fig.  16,  the  resonance  frequency  being  chosen  as  10,000 
cycles.  The  following  step-up  ratios  are  obtainable:  with  10,000-ohm  generator  (triode), 
1  to  4;  with  500-ohm  generator  (line),  1  to  17.8;  with  100-ohm  generator  (carbon  mike), 
1  to  40;  with  0.2-ohm  generator  (ribbon  mike),  1  to  890. 

PICK-TIP  AND  SHIELDING.  Input  and  interstage  transformers  often  work  at  very- 
low  voltage  levels.  As  a  result,  voltage  induced  in  the  windings  by  stray  magnetic  fields 
may  be  as  large  as  or  larger  than  the  signal  voltage.  These  stray  fields  are  produced  by 
nearby  power  transformers,  rectifier  filter  reactors,  open  loops  in  wires  carrying  large 
a-c  currents,  motor  generators,  etc.  A  hum  of  the  frequency  of  the  stray  field  is  introduced 
into  the  amplifier. 

Correction  of  hum  pick-up  should  begin  at  the  source,  if  possible.  Reduction  of  the 
flux  density  in  power  transformers  and  reactors,  by  proper  design,  placing  of  air  gaps  of 
reactors  inside  of  the  coils,  use  of  shielding  cans  around  power  transformers  and  reactors, 
and  tight  twisting  of  heavy-current  wiring  are  all  helpful. 

Removal  of  the  input,  or  interstage,  transformer  further  from  the  source  of  disturbance, 
and  orienting  it  so  that  its  coil  will  be  at  right  angles  with  the  coil  of  the  disturbing  trans- 
former, are  precautions  that  should  be  taken  when  laying  out  an  amplifier. 


6-22 


PASSIVE   CIRCUIT  ELEMENTS 


Input  transformers  are  sometimes  made  with  a  two-legged  core  (see  Section  3),  half 
of  the  primary  and  half  of  the  secondary  being  placed  on  each  leg.  The  two  halves  of 
each  winding  are  connected  in  series,  being  additive  for  flux  within  the  core,  but  subtractive 
for  external  fields.  Such  "hum-bucking"  construction  is  very  effective  in  reducing  hum 
pick-up,  if  the  external  field  is  uniform,  so  that  it  acts  upon  both  parts  of  the  transformer 
equally.  A  reduction  of  40  db  in  pick-up  may  be  realized. 

Shielding  of  the  input,  or  interstage,  transformer  is  also  very  effective  in  reducing  pick- 
up. A  drawn  nickel-alloy  case  with  a  tight-fitting  lid  will  reduce  pick-up  about  30  db. 
Two  such  nickel-alloy  shields,  one  inside  the  other,  separated  by  a  similar  copper  shield, 
will  reduce  pick-up  by  about  60  db.  Three  nickel-alloy  and  two  copper  shields  will  give 
about  90-db  reduction  of  pick-up.  Such  nested  shields  are  available.  See  E.  B.  Harrison, 
Notes  on  Transformer  Design,  Electronics,  February  1944. 


11.  DRIVER  TRANSFORMER 

Function.  The  Class  B  operated  output  stage  requires  an  auxiliary  stage  of  audio 
amplification  called  the  "driver"  stage.  The  driver  transformer  couples  the  plate  of  the 
driver  tube,  usually  a  triode,  to  the  grids  of  the  Class  B  amplifier.  The  function  of  the 
driver  stage  is  to  supply  to  the  grid  circuit  of  the  output  stage  large  positive  voltage  peaks, 
which  means  that  the  driver  stage  is  required  to  furnish  power.  In  this  respect  the  driver 
transformer  is  similar  to  the  output  transformer,  but  it  has  additional  requirements  im- 
posed upon  it  which  make  its  design  more  exacting. 

TURNS  RATIO.  The  secondary  load  on  a  driver  transformer  varies  over  a  wide 
range  during  each  half-cycle,  from  a  very  high  resistance  when  both  grids  are  negative  to  a 
low  resistance  when  either  grid  is  positive.  The  turns  ratio  of  the  transformer  must  be 
selected  so  that  the  change  in  load  resistance  has  a  negligible  effect  on  driver  tube  distor- 
tion, a  condition  which  is  satisfied  by  using  a  step-down  ratio.  The  value  of  the  turns 
ratio  is  a  compromise  between  distortion  and  driving  power. 

FREQUENCY  RESPONSE  is  governed  by  the  same  factors  as  for  the  output  trans- 
former, viz.,  primary  inductance  at  the  low  frequencies  and  leakage  inductance  at  the 
high  frequencies.  There  is  a  difference,  in  that  the  load  on  the  driver  transformer  is  not  a 
constant  resistance  but  varies  during  the  cycle.  Primary  inductance  should  be  high 
enough  to  give  the  desired  low-frequency  response  with  n,  =  °o.  Leakage  inductance 
should  be  low  enough  to  give  the  desired  response  when  TL  =  peak  grid  voltage  swing/peak 
grid  current  swing. 

LEAKAGE  AND  DISTORTION.  Leakage  reactance  in  the  driver  transformer  is  a 
reactance  in  series  with  the  grids  of  the  Class  B  stage.  It  causes  distortion  of  the  grid 
voltage  wave  at  the  higher  frequencies.  The  high-frequency  range  of  a  driver  transformer 

is  limited  by  distortion  rather  than  by  a  falling  off  of 
the  secondary  voltage.  Leakage  between  primary 
and  secondary  must  therefore  be  kept  to  the  mini- 
mum. It  is  also  important  that  each  half  of  the  sec- 
ondary be  coupled  equally  to  the  primary;  otherwise 
the  secondary  voltage,  at  high  frequencies,  will  not 
be  the  same  for  both  half-cycles,  which  will  produce 
even  harmonics  in  the  grid-voltage  wave.  The  re- 
quirements of  low  leakage  and  equal  coupling  are 
met  by  using  the  coil  arrangement  shown  in  Fig  17. 
See  T.  McLean,  An  Analysis  of  Distortion  in  Class 
B  Audio  Amplifiers,  Proc.  I.R.E.,  March  1936. 
Capacitances  of  the  windings  and  input  capacitance  of  the  tubes  have  very  little  effect 
upon  frequency  response  or  distortion.  However,  they  may  resonate  with  the  leakage 
reactance  at  some  superaudible  frequency  to  cause  parasitic  oscillations  of  the  Class  B 
stage.  Such  oscillations  cannot  be  suppressed  by  means  of  series  grid  resistors  without 
increasing  distortion.  It  is  desirable,  therefore,  to  shunt  a  small  capacitance  from  each 
Class  B  grid  to  ground. 


Fia.   17.    Winding  Arrangement,  Class 
B  Driver  Transformer 


12.  PHYSICAL  DESIGN  OF  AUDIO  TRANSFORMERS 

Data  Required.  From  the  foregoing  analyses  of  various  kinds  of  audio  transformers, 
it  is  evident  that  the  circuits  which  the  transformer  is  coupling  together  must  be  clearly 
and  completely  specified  before  a  design  of  the  transformer  is  attempted.  The  first  step 
in  designing  an  audio  transformer  is  to  draw  a  diagram  of  the  circuits,  showing  the  values 


PHYSICAL  DESIGN   OF  AUDIO  TRANSFORMERS 


6-23 


of  generator  and  load  impedance,  direct  current  in  any  winding,  and  other  pertinent  data. 
Next,  the  desired  constants  of  the  transformer,  such  as  turn§  ratio,  primary  inductance, 
leakage  inductance,  secondary  capacitance,  winding  resistances,  and  core-loss  resistance 
are  determined  from  the  required  performance  of  the  transformer.  Methods  of  finding 
what  constants  in  the  transformer  will  give  desired  performance  are  described  above. 

DESIGN  METHOD.  There  is  no  straightforward  method  of  going  at  the  physical 
design  of  an  audio  transformer.  The  design  is  carried  out  by  making  several  trials,  each 
one  approaching  closer  to  the  desired  constants.  Procedure  is  as  follows: 

1.  Assume  a  core  size  and  a  core  material.    As  a  starter,  a  core  of  El  scrapless  lamina- 
tions (see  Section  3,  Ferrous-cored  Inductors) ,  T/2  in.  to  3/4  in.  center  leg,  stack  equal  to 
width  of  center  leg,  and  silicon-steel  material,  might  be  chosen, 

2.  Calculate  the  number  of  primary  turns  that  will  give  the  desired  value  of  primary 
inductance  (Ferrous-cored  Inductors). 

3.  Multiply  the  number  of  primary  turns  by  the  turns  ratio  to  give  the  number  of 
secondary  turns. 

4.  Determine  the  primary  and  secondary  wire  sizes,  using  500  circular  mils  per  ampere 
as  a  first  trial,  but  not  using  wire  smaller  than  No.  41  AWG.    Extremely  small  and  light 
transformers  may  employ  wire  as  small  as  No.  44,  but  these  very  small  wire  sizes  should 
be  avoided,  if  possible,  because  of  breakage  when  winding.    If  direct  as  well  as  alternating 
current  is  present  in  a  winding,  the  total  current  rms  value  will  be 


Itotal  =  Vide2  +  /ac2 


(18) 


5.  Lay  out  the  windings.    As  a  rule  it  is  best  to  calculate  the  number  of  turns  per  layer 
and  the  number  of  layers  of  each  winding  rather  than  to  rely  on  some  winding  space 
factor.    The  arrangement  of  windings  may  be  fixed  by  some  special  requirements  of  the 
transformer,  as  shown  in  Figs.  13,  14,  and  17.     Unless  there  is  some  reason  for  doing 
otherwise,  the  primary  is  customarily  wound  first,  with  the  secondary  over  It.     (See 
Inductors  for  details  of  construction.) 

6.  If  the  total  calculated  build  of  the  coil,  including  spool,  layer  insulation,  and  wrapper, 
exceeds  90  per  cent  of  the  window  height,  the  core  is  too  small,  and  steps  1,  2,  3,  and  5 
must  be  repeated,  using  either  a  larger  core,  a  core  with  a  larger  window,  such  as  the  EE 
scrapless  style,  or  a  core  of  better  magnetic  material.    If  the  coil  build  is  far  below  90  per 
cent  of  the  window  height,  a  smaller  core  should  be  tried. 

7.  After  working  out  a  core  and  coil  that  will  fit  and  will  have  the  desired  primary 
inductance  and  turns  ratio,  the  other  constants  should  be  calculated  from  the  design. 
These  include  primary  and  secondary  resistances,  core-loss  resistance,  and  leakage  induct- 
ance.   In  the  case  of  input  and  interstage  transformers,  the  distributed  capacitance  and 
capacitance  to  ground  of  the  secondary  should  also  be  calculated.    In  the  case  of  high- 
level  output  and  modulation  transformers,  maximum  flux  density  at  the  lowest  frequency 
(see  Inductors),  middle-range  efficiency,  and  heating,  should  also  be  calculated. 

8.  Modify  the  design  as  required.    An  inspection  of  the  first  trial  design  will  suggest  the 
changes  needed.    Leakage  inductance  and  secondary  capacitance  can  be  varied  consider- 
ably by  changing  the  arrangement  and  shape  of  the  windings.    Resistance  of  the  windings 
can  be  changed  by  using  larger  or  smaller  wire. 

LEAKAGE  INDUCTANCE  depends  upon  coil  geometry.     For  a  two-winding  trans- 
former, as  shown  in  Fig.  18,  the  leakage  inductance,  referred  to  the  primary,  is 


Le 


I 


X  10~9     henry 


(19) 


in  which  c  =  length  of  a  mean  turn,  a  turn  halfway  between  the  innermost  and  outermost 
layers;  I  =  length  of  winding,  or  wire  traverse;  a  =  distance  between  windings,  copper 
to  copper;  di  and  dz  are  the  build-ups 

-I 


of  the  two  windings,  all  dimensions 
being  expressed  in  inches;  and  Np  is 
the  number  of  turns  of  the  primary 
winding.  The  method  of  deriving  this 
formula  is  given  by  R.  R.  Lawrence, 
Principles  of  A-C  Machinery.  Leakage 
inductance,  referred  to  the  secondary, 
is  given  by  the  same  formula,  except 
that  N/  is  used  in  place  of  Np*,  Ns  be- 
ing the  number  of  turns  of  the  second- 
ary winding. 


Core 

Secondary  Winding 


Primary  Winding 
Margin 


FIG.  18.    Cross-section  of  Transformer  Windings 


6-24 


PASSIVE  CIRCUIT  ELEMENTS 


Leakage  inductance  can  be  reduced  by  dividing  either  the  primary  or  the  secondary 
winding  into  two  sections,  placing  the  other  winding  between  the  two  sections.  One 
arrangement  is  shown  in  Fig.  19a,  in  which  the  windings  are  concentrically  wound.  A 
second  arrangement  is  shown  in  Fig.  196,  in  which  the  windings  are  coaxially  wound;  this 
construction  is  termed  "pancake"  winding.  For  the  two  cases: 


ScNJ 


heary 


(20a) 


(206) 


(o) 
FIG.  19.     Interleaved  Windings,    (a)  Concentrically  Wound.     (&)  Coaxially  Wound. 

DISTRIBUTED  CAPACITANCE  of  an  audio-frequency  transformer  winding  is  made 
up  of  the  layer-to-layer  capacitances.  The  turn-to-turn  capacitances  are  negligible.  The 
equivalent  capacitance  across  a  winding  is  the  resultant  of  the  layer-to-layer  capacitances, 
in  series.  A  winding  of  many  layers,  therefore,  has  less  distributed  capacitance  than  a 
winding  of  few  layers.  The  distributed  capacitance,  Cd,  is 


di* 


micro-microfarads 


(21) 


in  which  c  =  mean  length  of  turn  of  the  winding;  I  —  length  of  winding,  or  wire  traverse; 
d  =  distance  between  layers,  copper  to  copper,  all  dimensions  expressed  in  inches; 
T  =  number  of  layers  of  wire  in  the  winding;  and  k  =  average  dielectric  constant  of  layer 
insulation,  enamel,  and  impregnating  compounds.  For  paper-insulated  layers,  k  =  3, 
approximately.  See  J.  H.  Morecroft,  Principles  of  Radio  Communication,  2nd  Ed.,  pp. 
233-235. 

CAPACITANCE  TO  GROUND  or  between  windings  consists  of  the  capacitances  of  the 
inner  layer  and  of  the  outer  layer  of  a  winding  to  surfaces  which  are  adjacent  to  them. 
The  capacitances  at  the  ends  of  a  winding  are  usually  negligible.  Usually,  one  winding 
is  wound  over  another,  concentrically,  with  10  to  40  mils  of  insulation  between  them.  Also, 
the  wire  traverse  is  usually  the  same  for  both  windings.  The  capacitance  between  the  two 
windings  is  the  capacitance  between  two  parallel  surfaces,  of  the  same  area,  having  a  very 
small  separation  between  them. 

It  is  true  that  the  a-c  voltage  between  the  two  surfaces  is  not  usually  the  same  at  all 
points,  because  the  a-c  voltage  across  the  outer  layer  of  the  one  winding  is  usually  not  the 
same  as  that  across  the  inner  layer  of  the  other  winding.  Also,  the  capacitance  calculated 
between  the  outer  layer  of  the  one  winding  and  the  inner  layer  of  the  other  is  assumed  to 
be  from  the  finish  of  the  one  winding  to  the  start  of  the  other.  These  are  minor  errors 
if  the  number  of  layers  on  each  winding  is  greater  than  10. 

If  the  mean  circumference  of  the  space  between  the  two  windings  ~  c,  length  of 
winding  =  Z,  and  separation  of  windings  =  d,  all  dimensions  in  inches,  and  k  is  the  di- 
electric constant  of  the  insulation  between  them,  the  winding-to-winding  capacitance, 
Cwt  is 


a 


.  .      .       , 

micro-microfarads 


,00. 

(22) 


The  same  formula  may  be  used  to  compute  other  capacitances,  such  as  that  to  core 
to  shield. 


AUDIO  TRANSFORMER  MEASUREMENTS       6-25 


13.  AUDIO  TRANSFORMER  MEASUREMENTS 

RESISTANCE.     The  d-c  resistance  is  usually  accurate  enough  at  audio  frequencies. 

INDUCTANCE  AND  CAPACITANCE.  Most  iron-cored  transformers  for  audio  and 
power  frequencies  resonate  at  some  frequency  in  the  neighborhood  of  1000  cycles.  This 
is  a  parallel  resonance  of  the  mutual  inductance  and  the  winding  capacitances.  Measure- 
ment of  primary  or  secondary  inductance  or  winding  capacitance  at  1000  cycles  is  meaning- 
less. Primary  or  secondary  inductance  must  be  measured  at  some  low  frequency,  60 
cycles  being  a  convenient  one,  care  being  taken  to  apply  an  appropriate  value  of  a-c 
voltage  and  direct  current.  Bridges  for  inductance  measurement  are  described  in  Sec- 
tion 11. 

Capacitance  must  be  measured  at  some  high  frequency,  such  as  4000  cycles  or  above. 
Any  terminals  which  are  normally  at  a-c  ground  potential  should  be  grounded  during  such 
measurements.  A  measurement  of  capacitance  across  any  winding  will  include  reflected 
capacitances  from  other  windings.  If  it  is  desired  to  measure  the  secondary  capacitance 
of  an  input  or  interstage  transformer,  this  may  be  done  indirectly  by  measuring  the  leakage 
inductance  and  the  leakage  resonance  frequency. 

LEAKAGE  INDUCTANCE  is  most  conveniently  measured  on  a  1000-cycle  bridge, 
short-circuiting  one  winding  and  measuring  the  inductance  of  the  other.  This  frequency 
is  satisfactory  since  the  mutual  inductance  and  winding 

capacitances  are  shorted  out,  leaving  only  the  leakage  in-          I 

ductance  and  winding  resistances  as  factors  in  the  measure- 
ment.   If  the  leakage  inductance  referred  to  the  primary  is    100(J 
to  be  measured,  the  secondary  is  short-circuited  and  the  in-     oscillator* 
ductance  of  the  primary  is  measured.    See  Fig.  3,  in  which 
Es  would  be  short-circuited- 


TURNS   RATIO  is  most  accurately  and    conveniently 


measured  with  a  bridge  as  illustrated  in  Fig.  20.    Such  a      FIG.  20.    Turns-ratio  Bridge 
bridge  is  as  accurate  as  the  resistance  arms,  except  when 

there  is  a  large  amount  of  leakage  inductance  in  the  transformer.    Polarity  is  determined 
at  the  same  time  as  turns  ratio.    The  windings  must  be  additive  in  order  to  obtain  a  null. 


CORE-LOSS  RESISTANCE,  in  the  middle-frequency  range,  referred  to  the  primary, 
is  found  by  measuring  the  primary  impedance,  at  the  self-resonant  frequency  of  the 
transformer,  with  no  load  on  the  secondary.  Figure  21  shows  how  this  impedance  may 

be  measured.  This  impedance  is  practically  equal  to 
the  core-loss  resistance.  Referring  to  Fig.  3,  if  £Li, 
the  primary  inductance,  is  resonant  with  the  total 
capacitance  Ci  -}-  CYj  their  combined  impedance  is 

Osci  la  tor  JU-r  r-  -  ^e3[l|C  —       infinite,  and  they  have  no  shunting  effect  on  the  cir- 

I   pjj  I     prj.  oj  |K  Sec^    cuit.    If  rif  and  Cj/,  the  load  resistance  and  capaci- 
I      L          ylllC*  _       tance,  are  removed  from  the  transformer,  the  only 
—  I  -    >^    '  shunt  left  in  the  circuit  is  the  core-loss  resistance  rc. 

Vacuum-tube     The  series  elements  ri,  the  primary  winding  resistance, 

Voltmeter  _  Qf 


FIG,  21,  Measurement  of  Impedance  ance,  are  usually  very  small  as  compared  with  re  and 

can  be  neglected. 

FREQUENCY  CHARACTERISTIC.  An  audio  oscillator  and  a  suitable  voltmeter  are 
required.  For  measuring  input  or  interstage  transformers,  the  voltmeter  must  be  of  the 
tube  type,  but  for  output  transformers  thermocouple  or  rectifier-type  voltmeters  may  be 
used.  In  any  event,  the  voltmeter  should  not  appreciably  load  the  circuit  being  measured. 

It  is  essential,  when  measuring  frequency  response  or  phase  shift  of  an  audio  transformer, 
that  the  circuits  between  which  the  transformer  works,  on  the  primary  and  secondary 
sides,  be  either  included  in  the  measurement  or  that  equivalent  resistors,  capacitors,  etc., 
be  used.  A  measurement  of  the  transformer  alone  is  meaningless.  If  the  actual  tube  or 
line  on  the  primary  side  is  not  used,  an  equivalent  resistor  should  be  placed  in  series  with 
the  primary  winding. 

Measurements  should  be  made  at  an  a-c  voltage  level  corresponding  to  actual  operating 
conditions,  and  with  direct  current  in  the  windings  equal  to  any  unbalanced  direct  current 
under  actual  operating  conditions. 


6-26 


PASSIVE  CIRCUIT  ELEMENTS 


14.  POWER  TRANSFORMER 

The  function  of  the  power  transformer  in  radio  and  communication  equipment  is  three- 
fold: to  insulate  the  equipment  from  the  power  line,  to  reduce  the  line  voltage  to  the 
voltages  required  by  the  tube  heater  circuits,  and  to  step  up  the  line  voltage  to  energize 
the  anodes  of  the  rectifier  that  supplies  the  d-c  plate  and  bias  voltages.  The  secondary 
windings  that  supply  voltage  to  heater  circuits  are  usually  called  "filament"  windings,  and 
the  secondary  winding  that  supplies  voltage  to  rectifier  anodes  is  termed  the  "plate" 
winding,  A  typical  power  transformer  might  have  a  115-volt  60-cycle  primary,  a  600- volt 
center-tapped  plate  winding,  a  5-volt  2-ampere  filament  winding  for  the  rectifier  heater, 
and  a  6.3-volt  3-ampere  filament  winding  for  the  other  heaters.  Variations  of  this  basic 
pattern  are,  of  course,  very  numerous,  depending  upon  the  voltages  and  currents  needed. 
Additional  secondaries  may  be  required  to  supply  pilot  lamps,  relays,  control  motors,  etc. 
In  transmitters,  separate  rectifiers  may  be  used  to  supply  plate  and  bias  voltages,  requiring 
two  plate  windings.  It  may  be  desirable  to  turn  on  the  heaters  of  large  tubes  for  a  warm-up 
period  before  applying  the  plate  voltage,  which  means  that  two  separate  power  trans- 
formers are  required,  one  to  supply  the  heaters,  termed  a  filament  transformer,  and  one 
to  supply  the  rectifier  anodes,  termed  a  plate  transformer. 

Power-line  Frequency  is  60  cycles  per  second  throughout  most  of  the  United  States. 
In  some  parts  of  the  United  States  and  in  many  foreign  countries  50  cycles  is  standard. 
Most  power  transformers  for  radio  equipment  are  designed  for  50-  and  60-cycle  operation. 
In  aircraft,  400  or  800  cycles  per  second  is  often  used. 

VOLT-AMPERE  RATING.  The  volt-ampere  rating,  of  any  secondary  winding  is  the 
product  of  the  rms  voltage,  under  load,  by  the  rms  current.  The  total  volt-ampere  rating 
of  all  the  secondaries  is  the  sum  of  the  volt-ampere  ratings  of  the  individual  secondaries. 
For  a  filament  winding,  the  volt-ampere  rating  is  simply  a-c  voltage  times  a-c  current  (in 
amperes).  Both  voltage  and  current  are  of  sinusoidal  wave  form,  and  the  rms  value  of 
each  is  the  ordinary  a-c  effective  value.  For  a  plate  winding,  the  rms  voltage  is  the 
ordinary  a-c  voltage,  since  the  voltage  wave  is  sinusoidal.  However,  the  current  in  a  plate 
winding  is  not  sinusoidal.  Its  rms  value  depends  upon  the  amount  of  direct  current 
supplied  by  the  rectifier,  the  kind  of  rectifier  circuit  used,  and  whether  the  first  element 
of  the  rectifier  filter  is  an  inductor  (choke  input)  or  a  capacitor  (condenser  input).  Table  1 
gives  the  ratio  of  rms  current,  in  the  winding,  to  direct  current,  for  various  kinds  of  rec- 
tifier. The  factors  given  for  capacitor  input  are  round-number  approximations,  which 
are  accurate  enough  for  most  power-transformer  designs.  Actually  these  factors  for 
capacitor  input  vary  widely,  depending  upon  the  resistance  of  the  rectifier  and  the  resist- 
ance of  the  d-c  load.  For  analysis  of  rectifiers  see  O.  H.  Schade,  Analysis  of  Rectifier 
Operation,  Proc.  I.R.E.,  July  1943;  F.  E.  Terman,  Radio  Engineering,  2nd  Ed.,  pp.  479- 
500;  R.  W.  Armstrong,  Polyphase  Rectification  Special  Connections,  Proc,  I.R.E.,  Jan- 
uary 1931. 

Table  1.    Form  Factors  for  Plate-winding  Currents 


Type  of  Rectifier 

Heating  Current 

Volt-drop  Current 

Bridge,  inductor  input          

Irms/Idc 

1.0 

n 
1.0 

Full-  wave,  pushpull,  inductor  input  
Full-  wave,  pushpull,  capacitor  input.  .  .  . 
Bridge  capacitor  input       

0.707 
1.0 
1.5 

0.500 
0.707 
1   5 

2.0 

2  0 

Voltage-doubler,  capacitor  input  

3.0 

3.0 

SIZE  OF  POWER  TRANSFORMERS  is  governed  by  heating  rather  than  by  efficiency 
in  most  radio  and  communications  work.  Allowable  heating  limits  the  flux  density  in 
the  core  and  the  current  density  in  the  windings.  If  the  flux  density  is  fixed,  the  core 
cross-section  is  proportional  to  the  volts  per  turn.  See  eq.  (26),  below.  If  the 'current 
density  is  fixed,  the  core  window  area  is  proportional  to  the  ampere-turns. 

Core  cross-section  X  window  area  is  proportional  to  volts  per  turn  X  ampere-turns. 


A  X  W  -  p    ~  X 


p(E  X  /) 


(23) 


wiiere  A  is  the  cross-section  of  the  core,  W  is  the  area  of  the  core  window,  p  is  a  constant 
of  proportionality,  N  is  the  number  of  turns  on  the  secondary  of  a  transformer  having  but 
one  secondary,  E  and  I  are  the  rms  voltage  and  current  of  this  secondary,  and  E  X  /  is 
the  volt-ampere  rating  of  the  transformer.  When  there  are  several  secondaries,  the  same 


POWER  TRANSFORMERS 


6-27 


formula  holds,  but  E  X  I  is  the  sum  of  the  volt-ampere  ratings  of  all  the  secondaries. 
Applying  this  formula  to  the  common  case  of  a  60-cycle  power  transformer,  having  40 
to  50  deg  cent  temperature  rise,  dimensions  being  in  inches, 

A  X  W  =  0.024CE  X  I)     approximately  (24) 

Power  transformers  often  employ  laminations  of  the  El-scrapless  shape  (see  Section  3r 
Ferrous-cored  Inductors).  The  core  stack  is  often  equal,  or  nearly  so,  to  the  width  of 
the  center  leg;  that  is,  the  center  leg  has  a  square  cross-section.  This  core  shape  has  a 


0.1 


30 


40 


60  70  80 

Kilolines  per  Square  Inch 
FIG.  22.    Core  Loss.    Silicon  Steel  at  60  Cycles 


90 


100 


definite  relationship  between  window  area  and  core  cross-section,  W  =  0.75A.     Equa- 
tion (24),  for  this  case,  can  be  simplified  to 


approximately 


(25) 

These  formulas  are  fairly  accurate  for  the  average  run  of  power  transformers.  However, 
if  operating  voltages  are  high,  or  if  the  number  of  secondaries  exceeds  three,  a  larger 
than  usual  part  of  the  window  is  taken  up  by  insulation,  leaving  less  space  for  wire.  To 
allow  for  this,  a  somewhat  larger  core  size  should  be  chosen.  Often,  in  such  cases,  a  lami- 
nation shape  is  desirable,  having  more  window  area  than  the  El-scrapless  lamination, 
such  as  the  EE-scrapless  style. 

CONSTRUCTION  OF  POWER  TRANSFORMERS.  Compactness  is  accomplished 
by  the  use  of  a  shell-type  core,  using  laminations  of  the  El-scrapless  or  EE-scrapless 
shapes.  Laminations  are  stacked  alternately,  or  interleaved,  one  each  way  or  two  each 


6-28  PASSIVE    CIRCUIT  ELEMENTS 

way.  Core  material  is  almost  always  silicon  steel  having  2.5  per  cent  silicon  content,  or 
higher.  Lamination  thickness  may  be  14-miI  (U.  S.  gage  No.  29),  19-mil  (U.  S.  gage  No. 
26),  or  25-mil  (U.  S.  gage  No.  24).  The  choice  of  silicon  content  and  lamination  thickness 
is  a  compromise  between  cost  and  core  loss,  the  lower  silicon  content  and  greater  thickness 
having  the  higher  core  loss.  Core  loss  vs.  flux  density  for  several  typical  grades  and  thick- 
ness of  laminations  are  given  in  Fig.  22.  See  Ferrous-cored  Inductors,  for  lamination 
shapes,  kinds  of  core  material,  and  references. 

Coil  construction  usually  follows  a  conventional  pattern,  because  a  particular  routine 
of  winding  has  been  found  to  be  most  convenient.  A  single  coil,  which  consists  of  the 
various  windings,  is  made,  and  is  placed  upon  the  center  leg  of  the  shell-type  core.  The 
primary  is  wound  first,  over  a  formed  spool  of  paper  or  fiber.  The  winding  is  layer  wound, 
with  paper  insulation  between  layers.  Next,  an  electrostatic  shield,  consisting  of  one  turn 
of  thin  copper,  the  overlapping  ends  being  insulated  from  each  other,  is  placed  over  the 
primary.  The  high-voltage  or  plate  winding  is  wound  on  next.  Up  to  this  point,  the 
winding  is  done  in  multiple,  ten  or  more  coils  being  wound  at  a  time.  Generally,  at  this 
point,  the  coils  are  sawed  apart.  The  primary  and  plate  windings  ordinarily  use  wire  of 
sizes  too  small  to  be  brought  out  of  the  windings  as  leads.  So  flexible,  insulated  leads 
are  anchored  on  top  of  the  plate  winding,  the  wire  from  the  primary  and  plate  windings 
being  brought  around  the  ends  of  the  windings  and  soldered  to  these  leads.  Then  the 
filament  windings,  which  usually  consist  of  a  few  turns  of  large  wire,  are  wound  on  singly, 
either  one  over  the  other  or  side  by  side,  depending  on  the  number  of  turns,  wire  size,  and 
wire  traverse  required.  Generally  the  wires  used  for  filament  windings  are  sufficiently 
large  and  rugged  to  be  extended  out  of  the  windings  as  leads.  See  Ferrous-cored  Inductors, 
and  also  H.  C.  Roters,  Electromagnetic  Devices,  Chapter  VI. 

The  coil,  or  the  core  and  coil  together,  are  baked  dry  and  impregnated  with  varnish, 
wax,  or  asphaltic  compound  to  exclude  moisture,  strengthen  the  coil  mechanically,  and 
reduce  lamination  hum. 

DESIGN  PROCEDURE  is  relatively  straightforward  for  power  transformers,  so  that 
design  calculations  are  often  made  on  a  standard  form  sheet,  or  calculation  sheet.  Steps 
are  as  follows: 

1.  Determine  the  rms  voltages  and  currents  of  the  secondary  windings  and  the  total 
volt-ampere  rating  of  all  the  secondaries. 

2.  Choose  a  core  size  which  will  satisfy  eqs.  (24)  or  (25). 

3.  Determine,  approximately,  the  primary  current.    Assume  90  per  cent  efficiency  and 
90  per  cent  power  factor  as  a  first  approximation.     If  the  plate  winding  is  not  center- 
tapped,  the  volt-ampere  input  to  the  primary  will  be 

(Filament  volt-amps  +  Plate  volt-amps)  -4-  0.81 

If  the  plate  winding  is  center-tapped,  the  volt-ampere  input  to  the  primary  will  be 
(Filament  volt-amps  -f  0,707  Plate  volt-amps)  -r-  0.81 

4.  Calculate  primary  turns,  Np. 

(26) 

in  which  EP  —  primary  voltage,  /  =  frequency,  B  =  flux  density  in  lines  per  square  inch, 
A  —  cross-sectional  area  of  core  in  square  inches,  and  k  =  stacking  factor.  Flux  density 
is  usually  around  70  kilolines  per  square  inch,  at  the  nominal  primary  voltage,  for  60-cycle 
designs. 

5.  Calculate  secondary  turns.    Assuming  a  regulation  of  10  per  cent,  as  a  first  trial, 

Ns  =  ^  X  Np  X  1.10  (27) 

tip 

If  the  number  of  turns  on  the  low-voltage  filament  windings  comes  out  fractional,  the 
turns  on  all  windings  should  be  increased  or  decreased  sufficiently  to  give  each  low-voltage 
winding  a  whole  number  of  turns. 

6.  Determine  wire  sizes  of  all  windings.    The  rule  of  1000  circular  mils  per  ampere  is 
very  convenient  to  use  as  a  first  approximation  and  is  usually  not  far  from  the  correct 
value  arrived  at  in  the  final  design. 

7.  Lay  -out  the  windings.    This  is  done  by  calculating  the  number  of  turns  per  layer, 
the  number  of  layers,  and  the  build  of  each  winding.    See  above,  "Construction  of  Power 
Transformers,"  and  Ferrous-cored  Inductors.    Margins  are  usually  1/8  in.    It  is  considered 
good  practice  to  use  a  double  thickness  of  layer  insulation  between  the  two  inside  layers 
and  between  the  two  outside  layers  of  the  high-voltage  winding,  to  provide  a  factor  of 
safety  against  transients  caused  by  line  surges  or  switching.    The  total  build  of  the  coil, 


POWER  TRANSFORMERS  6-29 

including  spool,  insulation,  and  outside  wrapper,  should  not  exceed  90  per  cent  of  the  win- 
dow height. 

Insulation  between  windings  is  determined  by  the  operating  voltage.  The  rule  of  twice 
normal  plus  1000  volts,  for  test  voltage,  is  commonly  used.  Insulation  must,  of  course, 
be  able  to  stand  something  more  than  the  test  voltage.  The  factor  of  safety  allowed  is 
governed  by  cost  and  by  the  type  of  service  which  the  radio  or  communications  equipment 
is  required  to  give.  For  test  voltages  of  2500  rms  or  less,  several  thicknesses  of  Kraft 
paper  or  two  turns  of  varnished  cambric  are  enough  insulation  between  windings.  Margins 
of  Vs  in.  are  adequate  for  creepage.  Above  2500  volts  rms  test,  wider  margins  are  neces- 
sary as  well  as  better  insulation  between  windings.  If,  for  example,  the  test  voltage  is 
4000,  and  a  two-to-one  safety  factor  is  allowed  in  the  design,  the  margins  must  be  wide 
enough  to  withstand  8000  volts  rms  creepage.  This  requires  about  7/i6-in.  margins.  A 
point  is  reached  where  the  margins  required  for  creepage  use  up  too  much  of  the  available 
winding  space.  Such  windings  are  insulated  by  wrapping  them  with  half-lapped  tape,  of 
varnished  cambric,  Fiberglas,  or  other  insulating  material.  A  narrow-windowed  lamina- 
tion or  side-by-side  sections  of  the  winding  are  employed  with  very  high  voltage  coils,  of 
1000  volts  or  higher,  in  order  to  keep  the  turns  per  layer  and  the  voltage  per  layer  low. 

CALCULATION  OF  PERFORMANCE.  The  procedure  outlined  above  will  give  a 
design  which  is  usually  not  very  far  from  a  satisfactory  final  design.  It  is  necessary, 
however,  to  calculate  the  output  voltages  of  the  secondaries,  the  heating,  regulation,  and 
efficiency,  and  then  to  make  minor  adjustments  in  the  design  as  required. 

1.  Resistance  of  each  winding  is  calculated  from  the  mean  length  of  turn,  in  feet,  the 
number  of  turns,  and  the  resistance  of  copper  wire,  of  the  particular  size,  in  ohms  per 
1000  ft  (see  Section  2,  for  tables).  The  d-c  resistance  of  the  wire  is  used.  An  allowance 
should  be  made  for  the  fact  that  ordinary  power  transformer  windings  operate  at  tempera- 
tures higher  than  20  deg  cent,  usually  in  the  neighborhood  of  SO  deg  cent.  The  resistance 
of  copper  wire  increases  about  0.4  per  cent  per  degree  centigrade  above  20  deg  cent.  The 
mean  length  of  turn,  m,  in  feet,  is  given  by 

2(X  +  7)  -  SR  +  7r(2R  +  &) 

(28) 


in  which  X  and  T  =  inside  dimensions  of  the  winding,  R  =  inside  corner  radius,  and 
6  —  build  of  the  winding,  all  dimensions  being  expressed  in  inches. 

2.  Copper  loss  of  each  winding  is  the  rms  current  squared  X  resistance  of  that  winding. 
Total  copper  loss  is  the  sum  of  the  copper  losses  of  the  individual  windings. 

3.  Core  loss  is  the  product  of  the  watts  per  pound  (Fig.  22)  times  the  weight  of  the 
core  in  pounds.    Flux  density,  for  determining  watts  per  pound,  is  calculated  from  eq.  (26)  . 
Actually,  the  flux  density,  under  load,  is  somewhat  less  than  given  by  this  equation,  owing 
to  voltage  drop  in  the  primary  winding,  but  the  error  is  very  small.    See  Magnetic  Circuits 
and  Transformers,  Chapter  V. 

4.  Heating  may  be  calculated  from  the  losses.    There  is  no  accurate  formula  for  heating 
of  a  power  transformer  in  radio  or  communications  equipment,  because  so  many  unpre- 
dictable factors  are  involved.    The  proximity  of  other  hot  objects,  such  as  rectifier  tubes 
or  bleeder  resistors,  the  amount  of  ventilation,  the  nature  of  the  surfaces  toward  which 
the  transformer  is  radiating  heat,  whether  dull  or  shiny,  the  kind  of  finish  on  the  outside 
of  the  transformer,  and  the  characteristics  of  the  potting  compound,  if  the  transformer  is 
potted,  all  influence  the  temperature  rise  of  the  transformer.    However,  a  rough  rule  can 
be  worked  out,  based  upon  the  Stefan-Boltzmann  law  for  radiation  from  a  black  body, 
which  gives  an  approximate  figure  for  temperature  rise  and  is  valuable  as  a  basis  for 
comparing  one  transformer  design  against  another.    The  radiation  surface  of  a  core  and 
coil  is  only  slightly  more  than  the  total  surface  of  the  core,  including  window  area.    That 
is,  the  coil  surface  adds  very  little  to  the  radiation  surface  of  the  core  and  coil.    E.g.,  if 
the  outer  dimensions  of  a  laminated  core  are  2  1/2  by  3  in.  and  the  stack  is  1  in.,  the  total 
core  surface  is  26  sq.  in.    This  surface  is  very  easy  to  compute  from  the  core  dimensions 
and  may  be  taken  as  the  radiating  surface  of  the  core  and  coil.    Calling  this  surface  A, 
in  square  inches,  and  the  total  watts  loss  W,  the  temperature  rise  in  degrees  centigrade, 
AT,  is  roughly  given  by 

(29) 


See  Magnetic  Circuits  and  Transformers,  Chapter  VIII;  Thermal  Characteristics  of  Trans- 
formers, V.  M.  Montsinger,  Gen.  Elec.  Rev.,  April  1946;  and  R.  Lee,  Electronic  Trans- 
formers and  Circuits,  Wiley,  pp.  37-44. 

5.  Output  voltages  of  the  secondaries  under  load,  and  the  regulation  of  a  transformer, 
are  calculated  from  tho  resistances  of  the  various  windings.    Voltage  drop  in  a  transformer, 


6-30  PASSIVE   CIRCUIT  ELEMENTS 

due  to  leakage  reactance,  is  very  small  at  60  cycles  and  is  usually  not  worth  the  trouble  of 
computing.  An  allowance  of  1  per  cent  leakage-reactance  drop  is  usually  accurate  enough. 
The  full-load  voltage  of  the  various  secondary  windings  is  obtained  as  follows  : 

A.  Primary  per  cent  IB,  drop  is 

Pr,  %  =  J,X*X100  (30) 

&p 

in  which  Ip  =  in-phase  component  of  primary  current  =  rms  primary  current  -s-  power 
factor,  ri  —  resistance  of  primary  winding,  Ep  =  impressed  primary  voltage. 

B.  No-load  secondary  voltages  =  impressed  primary  voltage  X  turns  ratio,  for  the 
various  secondary  windings. 

C.  High-voltage  winding,  full-load  voltage  is 


X  0.99  (31) 


in  which  EQ  =  no-load  voltage  of  winding,  lac  =  direct-current  rectifier  load,  n  =  factor 
depending  on  type  of  rectifier,  and  r%  —  resistance  of  high-voltage  winding.  The  0.99 
multiplier  is  an  allowance  for  leakage-reactance  drop.  Values  of  the  n  factor  are  given 
in  Table  1.  To  illustrate  where  the  n  factor  comes  from,  consider  a  full-wave,  inductor- 
input  rectifier  with  winding  center-tapped.  The  direct  current  flows  in  one  half  of  the 
plate  winding  at  a  time,  so  the  voltage  drop  in  the  plate  winding  is  0.5  X  r%  X  /dc-  The 
value  of  n  is  0.5. 

D.  Filament  winding,  full-load  voltage  is 


X  0.99  (32) 


in  which  EQ  —  no-load  voltage  of  the  filament  winding,  1$  =  rms  load  current,  ra  =  resist- 
ance of  filament  winding. 

The  regulation  of  each  secondary  winding,  expressed  in  per  cent,  is 

No-load  volts  —  Full-load  volts 

TT      7i     l  1  1  X      -LUU 

Full-load  volts 

Ordinarily,  regulation  is  between  5  and  10  per  cent. 

6.  Efficiency  is  the  ratio,  output  watts  -f-  input  watts.  The  output  wattage  of  any 
filament  winding  is  simply  the  product  of  a-c  voltage,  under  load,  by  a-c  load  current. 
The  output  wattage  of  a  plate  winding  is  a  complicated  product  of  a  sinusoidal  voltage 
and  a  non-sinusoidal  current.  As  an  approximation,  multiply  the  rms  voltage  of  the 
total  plate  winding  by  the  volt-drop  current  as  given  in  Table  1.  Then  the  output  wattage 
of  the  transformer  is  the  sum  of  the  wattages  of  the  various  secondaries. 

Input  wattage  is  computed  by  multiplying  the  no-load  voltage  of  each  secondary  by  its 
rms  load  current  if  a  filament  winding,  or  by  the  volt-drop  current  if  a  plate  winding,  and 
adding  the  core-loss  watts.  The  efficiency  of  a  typical  power  transformer  is  about  90 
per  cent. 

16.  VIBRATOR  TRANSFORMER 

Function.  The  vibrator  transformer  is  used  in  radio  and  communications  equipment 
when  the  source  of  power  is  a  battery  instead  of  an  a-c  line.  This  occurs  in  mobile  applica- 
tions, such  as  automobile,  aircraft,  and  railroad.  The  vibrator  transformer  takes  the  place 
of  the  usual  power  transformer,  supplying  the  proper  voltages  to  rectifier  anodes  and  to 
heaters. 

The  center  tap  of  the  primary  winding  is  connected  permanently  to  one  end  of  the 
battery.  The  other  end  of  the  battery  is  connected  alternately  to  one  end  or  the  other 
of  the  primary  winding  by  means  of  contacts  on  a  vibrating  reed.  The  frequency  of  the 
reed  is  usually  between  100  and  200  cycles  per  second,  although  frequencies  as  high  as 
400  cycles  are  used.  This  applies  an  alternating  voltage,  of  square  wave  form,  to  the  pri- 
mary of  the  transformer.  The  frequency  of  this  alternating  voltage  is,  of  course,  the  same 
as  the  vibration  frequency  of  the  reed.  The  transformer  is  able  to  step  this  primary  voltage 
up  or  down,  as  required,  in  much  the  same  manner  as  an  ordinary  a-c  voltage. 

An  appreciable  time  is  required  for  the  reed  to  move  from  one  side  to  the  other,  so  that 
the  contacts  are  closed,  one  way  or  the  other,  only  about  80  per  cent  of  the  time.  The 
transformer  is  connected  across  the  battery  only  that  percentage  of  the  time.  The  portion 
of  the  time  that  the  vibrator  contacts  are  closed  is  called  the  time  efficiency  of  the  vibrator. 


VIBRATOR  TRANSFORMER  6-31 

A  capacitor,  called  a  "buffer"  capacitor,  must  be  connected  across  either  the  primary 
or  the  secondary  winding.  Usually  it  is  placed  across  the  highest  voltage  winding  because 
a  smaller  value  of  capacitance  will  suffice  there.  The  buffer  capacitor  is  an  essential  part 
of  the  transformer.  When  the  correct  value  of  buffer  is  used,  it  gives  a  smooth  cross-over 
^of  the  transformer  voltage  during  the  time  interval  when  the  vibrator  contacts  are  open. 
In  so  doing,  it  prevents  sparking  and  high-frequency  transients  at  the  break  and  at  the 
make. 

Often,  heater  circuits  are  connected  directly  to  the  battery,  leaving  only  rectifier  anode 
voltage  to  be  supplied  by  the  vibrator  transformer.     The  rectifier  winding  may  be  con- 
nected to  vacuum-tube  rectifiers  in  the  same 
manner  as  the  plate  winding  of  an  ordinary  [   j          ______  ^nfi   J_Buffer  Capacitor 

power  transformer.    Or  rectification  may  be 

accomplished  by  using  a  second  pair  of  con- 

tacts on  the  vibrator  which  switch  the  load 

back  and  forth  across  the  secondary  in  syn- 

chronism with  the  switching  of  the  battery 

across  the  primary.     Figure  23   shows  the 

circuit    of    a    synchronous    vibrator    trans-         FIG.  23.    Circuit  of  Synchronous  Vibrator 

former. 

DESIGN.  (Rectifier  secondary  only.)  A.  Core  size  is  much  larger  than  for  an  ordi- 
nary power  transformer  of  the  same  frequency  and  giving  the  same  d-c  voltage  and 
current.  As  battery  voltage  varies  widely,  flux  density  must  be  kept  low.  The  primary 
winding  is  almost  twice  as  bulky  as  that  of  an  ordinary  transformer.  The  current  is 
flowing  only  part  of  the  time,  and  so  larger  wire  is  required  for  the  same  average  current. 
On  the  other  hand,  if  the  frequency  of  the  vibrator  is  much  higher  than  60  cycles,  e.g., 
150  cycles,  the  higher  frequency  will  reduce  the  size  of  the  vibrator  transformer.  It  will 
still  be  somewhat  larger  than  an  equivalent  60-cycle  power  transformer. 

B.  Flux  density  and  primary  turns.  The  flux  density  in  a  vibrator-transformer  core 
is  given  by 

S  per  s<3uare  ^-^  (33) 


in  which  EI  =  normal  voltage  of  the  battery,  p  —  time  efficiency,  Np  —  total  primary 
turns,  A  —  cross-section  of  core  in  square  inches,  k  =  stacking  factor  of  core,  and  /  =  vi- 
brator frequency.  Flux  density  should  be  kept  below  40,000  lines  per  square  inch  at 
normal  battery  voltage.  A  still  lower  figure  may  be  desirable  at  higher  frequencies  because 
of  core  loss. 

C.  Secondary  turns.    As  a  first  approximation,  use 


Ns  =  1.33  X  Np  X  ——d-c  output  v^0 
p  Battery  volts 

D.  Current  and  wire  size.    The  rms  current  in  the  secondary  winding  is 

-  (35a) 

Vp 
The  rms  current  in  the  primary  winding  is 

r        0.707  X  /dc  ^   N%  /0,,, 

J-  p   —    7^= X   ~TT~  U>OO) 

VP         NP 

in  which  JdC  —  d-c  load  current  of  the  secondary.  Wire  sizes  should  be  chosen  to  have 
about  800  circular  mils  per  ampere.  Usually  the  primary  has  relatively  few  turns  because 
the  battery  voltage  is  a  low  voltage.  The  primary  wire  size  should  be  chosen  so  as  to  give 
2,  4,  6,  or  8  even  layers.  This  will  place  the  center  tap  at  the  end  of  a  layer,  which  is  con- 
venient for  multiple  winding. 

E.  Output  voltage.     The  total  resistance  of  the  transformer  and  associated  circuits, 
referred  to  the  working  half  of  the  secondary,  is 

TT~  \2  +  r**TL)  \N~P)  +2  +Ts    0hmS  (*36) 

in  which  the  r  symbols  indicate  resistances,  as  follows:  n  -  total  primary,  rv  —  vibrator 
contact,  TL  =  battery  leads,  rz  =  total  secondary,  and  rg  =  rectifier  tube,  if  any. 
During  the  time  that  the  vibrator  contacts  are  closed,  the  secondary  current 
Then,  the  IR  drop  expressed  in  volts  on  the  secondary  is  TT  X 


6-32  PASSIVE   CIRCUIT  ELEMENTS 

The  no-load  d-c  voltage  across  the  first  filter  capacitor  is 

Eo  =  aE1  jjf  (37) 

jyp 

a  being  a  constant  which  takes  care  of  imperfect  contacts,  primary  and  secondary  contacts 
not  exactly  synchronized,  etc.,  =  0.94  for  synchronous  rectifier,  0.98  for  tube  rectifier. 
Then  the  full-load  d-c  voltage  across  the  first  filter  capacitor  is 

(38) 

See  T.  T.  Short  and  J.  P.  Coughlin,  Try  the  Inverter  Transformer,  Mec.  Mfg.,  June  1946; 
F.  E.  Terman,  Radio  Engineering,  pp.  500-501;  Mallory  Vibrator  Data  Book. 

16.  PULSE  TRANSFORMER 

In  radar  work,  iron-cored  transformers  are  used,  which  have  voltage  impressed  upon 
them  for  very  short  periods.  These  pulses  are  repeated  at  regular  intervals,  the  time 
interval  between  pulses  being  perhaps  1000  times  the  pulse  duration.  Analysis  of  such 
pulse  transformers  cannot  be  made  on  a  basis  of  Fourier  analysis  of  the  wave  form  because 
of  the  relatively  great  time  interval  between  pulses.  Each  pulse  is  a  separate  transient, 
the  effect  of  which  dies  out  before  the  next  pulse,  except  for  core  magnetization. 

If  a  voltage  is  applied  suddenly  to  the  primary,  through  a  generator  resistance  such  as 
the  plate  resistance  of  a  modulator  tube,  then  is  held  constant  for  a  short  time  interval, 
and  then  is  suddenly  removed,  the  input  voltage  will  be  of  square  pulse  shape.  The  trans- 
former will  step  up,  or  step  down,  this  voltage,  in  accordance  with  its  turns  ratio,  but  the 
output  voltage  will  not  faithfully  follow  the  square  pulse  shape.  An  appreciable  time  is 
required  for  the  secondary  voltage  to  build  up  from  zero  to  its  maximum  value.  This 
"rise  time"  is  caused  by  the  leakage  inductance  and  capacitance  of  the  transformer.  Dur- 
ing the  time  that  the  input  voltage  is  being  held  constant,  the  output  voltage  will  be 
falling  off,  the  amount  of  drop  from  a  constant  voltage  value  being  inversely  proportional 
to  the  primary  inductance.  When  the  input  voltage  is  removed,  the  secondary  voltage 

does  not  drop  instantly  to  zero  but  drags  out 
-Input  Pulse  through  several  damped  oscillations  caused 

by  the  discharge  of  magnetic  energy  stored 
in  the  core  through  the  winding  capacitances 
and  load  resistance.     Figure  24  illustrates 
the  kind  of  deformation  of  a  pulse  that  oc- 
curs when  it  is  passed  through  a  transformer. 
See  R.  Lee,  Iron-core  Components  in  Pulse 
FIG.  24.    Deforming  of  Pulse  Shape  by  Trans-     Amplifiers     Dromes,    August    1943     and 
former  R-    Lee,    Electronic    Transformers    and    Cir- 

cuits, Wiley,  Chapter  IX. 

The  pulse  transformer  has  a  number  of  unique  features.  Because  of  the  very  low-duty 
cycle,  tremendous  pulse  power  can  be  handled  by  a  very  small  transformer.  The  require- 
ment of  low  leakage  inductance  is  met  by  close  spacing  between  the  primary  and  secondary 
windings  and  by  making  the  primary  layer  or  layers  exactly  the  same  length  as  those  of 
the  secondary,  even  to  the  extent  of  winding  several  wires  in  parallel  on  the  low-voltage 
winding.  If  a  winding  requires  more  than  one  layer,  the  conventional  forward-and-back 
method  of  winding  is  not  used;  layers  are  all  started  from  the  same  end  to  minimize 
capacitance  between  layers. 

The  small  physical  size  and  the  relatively  few  turns  which  are  essential  to  obtain  low 
leakage  inductance  together  with  very  high  pulse  voltages  make  voltage  gradients  neces- 
sary that  are  unheard  of  in  ordinary  transformer  design.  For  example,  200  or  more  volts 
per  turn  is  not  uncommon.  This  is  accomplished  by  using  Formvar-coated  wire  and  im- 
pregnating the  transformer  with  transformer  oil  under  a  very  high  vacuum  to  remove 
every  trace  of  air. 

High  primary  inductance  is  required,  to  keep  the  drop  of  voltage  during  the  pulse  to 
the  minimum.  A  new  conception  of  incremental  permeability  is  necessary  in  calculating 
the  inductance.  All  the  pulses  are  of  the  same  polarity;  consequently  the  core  is  left  in  a 
partially  magnetized  condition,  owing  to  remanence.  The  hysteresis  loop  described  by 
the  core  has  this  remanent  point  as  one  of  its  ends.  The  shape  of  the  loop  is  largely  con- 
trolled by  eddy-current  loss  when  the  pulses  are  very  short,  such  as  of  1  microsecond 
duration.  Under  these  conditions  incremental  permeability  is  much  lower  than  when  a 


INTRODUCTION  6-33* 

core  is  operated  with  alternating  current,  and  core  material  having  low  eddy-current  loss 
is  desirable.  Ribbon  cores  of  very  thin  silicon-steel  ribbon,  1  to  3  mils  thick,  are  very 
satisfactory  for  pulse  transformer  cores.  Their  incremental  permeability  under  micro- 
second pulse  conditions  is  about  300. 


ELECTRIC  WAVE  FILTERS 

By  A.  J.  Grossman 

A  wave  filter  is  a  device  for  separating  waves  characterized  by  a  difference  in  frequency. 
The  general  purpose  of  an  electric  wave  filter  is  to  separate  sinusoidal  electrical  currents 
of  different  frequencies.  Ideally,  a  filter  transmits  freely  the  currents  of  all  frequencies 
lying  within  a  specified  range  and  excludes  currents  of  all  other  frequencies.  It  may  be 
used  to  transmit  the  intelligence  contained  in  a  certain  band  and  exclude  adjacent  steady- 
state  interference;  combinations  of  filters  may  divide  a  wide  frequency  band  into  a  number 
of  relatively  narrow  channels  or  may  direct  selected  bands  from  one  transmission  path 
into  two  or  more  different  paths.  Except  in  the  simplest  forms,  a  filter  is  a  composite 
network  made  up  of  several  sections  connected  in  tandem.  Each  section  consists  of 
simple  arrangements  of  two-terminal  reactance  networks.  These  reactances  are  provided 
by  combinations  of  ordinary  coils  and  condensers,  crystals,  coaxial  lines,  and/or  wave 
guides. 

The  point  of  view  developed  by  Bode,  Campbell,  Cauer,  Foster,  and  others  for  the 
analysis  of  a  network  is  to  regard  the  combination  of  inductances  and  capacitances  as  a 
system  excited  by  a  vibratory  disturbance  to  which  the  methods  of  particle  dynamics  can 
be  applied.  It  is  convenient  to  express  the  analysis  in  terms  derived  from  the  classical 
theory  of  wave  propagation  in  continuous  media.  These  terms  are  the  image  impedance 
and  the  image  transfer  constant.  In  this  terminology  a  filter  is  described  as  a  system 
having  the  following  idealized  properties.  Signals  lying  within  a  preassigned  frequency 
band  are  transmitted  without  reduction  in  amplitude.  This  band  is  bounded  by  cutoff 
frequencies  at  which  there  is  an  abrupt  transition  from  free  transmission  to  attenuation. 
The  attenuation  increases  more  or  less  rapidly  with  frequency  as  the  departure  from  a 
cutoff  increases.  Concomitantly,  the  impedance  is  a  pure  resistance  in  the  pass  band  and 
changes  abruptly  at  a  cutoff  to  a  pure  reactance. 

These  idealized  characteristics  are  approached  in  an  actual  filter  inserted  between 
resistance  terminations.  The  insertion  loss  for  frequencies  in  the  pass  band  remote  from 
the  cutoff  is  essentially  nil  (apart  from  the  effect  due  to  dissipation  in  the  components). 
As  the  cutoff  is  approached  the  loss  increases.  The  transition  from  the  theoretical  pass 
band  to  the  attenuating  band  is  smooth.  This  transition  interval  can  be  made  extremely 
narrow  in  an  elaborate  design.  The  cutoff  is  not  a  frequency  at  which  there  is  an  abrupt 
change  from  zero  attenuation  to  a  large  value;  it  may  mark  the  point  at  which  there  is  a 
rapid  change  from  a  small  to  a  large  value  of  attenuation,  but  at  a  finite  rate.  This  de- 
parture from  the  idealized  characteristics  arises  from  the  fact  that  the  filter  is  not  termi- 
nated in  its  image  impedances.  In  simple  cases,  the  image  impedance  varies  considerably 
with  frequency,  and,  consequently,  the  input  impedance  of  the  filter  has  a  non-constant 
resistance  component  and  an  associated  reactance  component.  By  careful  design  the 
image  impedance  may  be  maintained  nearly  constant  over  almost  all  of  the  pass  band. 
Nevertheless,  there  is  still  the  smooth,  but  rapid,  transition  of  the  input  impedance  from  a 
predominantly  resistive  characteristic  in  the  pass  band  to  a  predominantly  reactive  char- 
acteristic  in  the  attenuating  band.  The  usual  types  of  filters  are  low-pass,  high-pass,  and 
band-pass. 

17.  INTRODTJCTION 

A  general  filter  network  may  be  represented  by  a  box,  as  in  Fig.  1,  having  two  pairs  of 
accessible  terminals.  The  performance  of  this  network  is  described  in  terms  of  its  image 
impedances  and  image  transfer  constant.  The  image  imped- 
ances are  defined  as  those  impedances  with  which  the  network  1  o- 
must  be  terminated  so  that  there  will  not  be  reflections  at  the  2  j  x- 

junctions  1—1'  and  2-2'.  That  is,  when  the  terminating  im-     lro         

pedance  £7 -is  connected  across  1-1',  then  the  impedance   FIG_  L    A  General  p^  Net_ 
measured  at  the  2-2'  terminals  is  Zj  •  and,  similarly,  when  work 

Zi^  is  connected  across  2-2',  the  impedance  measured  at  1-1' 

is  Zir     The  image  transfer  constant,  0,  is  defined  to  be  equal  to  one-half  the  natural  loga- 
rithm of  the  ratio  of  the  volt-amperes  flowing  into  the  network  to  the  volt-amperes  flowing 


6-34 


PASSIVE   CIRCUIT  ELEMENTS 


ntege: 


r~bO( 
HH 


HH 

C 


'2 


-[( 


Lzk+l  '• 


Ol  =  1 .  [•***••• -~.-[' 

ti       Lcoicos  •  •  •  O)n_i-J 


z  = 


Jco(w22  —   co2)  (co42  —  w2)    •  •  •  (con-1     ~  w  ) 


4-P) I 


=cx      =3 


•"[(•^-^zlw*"''2-'"'^ 


'-[( 


-Z^2A+1   = 


052034   •••ton_ 


3 


FIG.  2.    Design  Information  for 

out  of  the  network  when  the  network  is  terminated  in  its  image  impedances.  These  quanti- 
ties are  an  exact  measure  of  the  performance  of  the  filter  only  if  the  actual  terminations 
are  equal  to  the  image  impedances.  In  a  practical  design,  account  must  be  taken  of  reflec- 
tion and  interaction  effects  arising  from  the  mismatch  between  the  terminating  impedances 
of  the  filter  and  its  image  impedances.  These  effects  may  be  evaluated  by  the  method 
described  in  Section  5.  They  will  not  be  considered  in  detail  here.  It  will  be  assumed 
that  the  performance  of  the  filter  is  described  by  its  image  transfer  constant. 
By  writing  the  mesh  or  nodal  equations  for  the  network,  it  may  be  shown  that: 

(1) 
(2) 

(3) 

where  ZA  is  the  impedance  measured  at  the  1-1'  terminals  when  a  short  circuit  is  placed 
across  the  2-2'  terminals,  and  ZQI  is  the  impedance  measured  at  the  1-1'  terminals  when 
the  2-2'  terminals  are  open.  The  short-  and  open-circuit  impedances  Z&  and  ZQ%  at  the 
2-2'  terminals  are  measured  similarly. 


INTRODUCTION 


6-35 


(3) 


Z  =  3- 


;     n  =  odd  integer 


HP 


Ln-2 
P^"1       Ln 

pTRJtf> — o 

LjlJ 
C«.i 


r ^ i       ; 

I— (o>2Jfc— 1     —   Ur)Z  J&>"<<>2jfc— 1 


1,2,  •- • 


L!  =  B  - 


z-y- 


n_22  -  a:2)   . 


£2  £4 


" 


'2 


Two-terminal  Reactive  Networks 

The  short-  and  open-circuit  impedances  are  driving  point  impedances  of  a  purely  re- 
active network.  The  requirements  on  such  an  impedance  are  (by  Foster's  theorem) : 

It  is  an  odd  rational  function  of  the  frequency,  OJ/STT,  which  is  completely  determined, 
except  for  a  constant  factor,  #,  by  assigning  the  resonant  and  antiresonant  frequencies, 
subject  to  the  condition  that  they  alternate  and  include  both  zero  and  infinity.  Such  an 
impedance  function  may  be  physically  realized  by  several  canonical  structures,  among 
them  a  combination  of  antiresonant  circuits  connected  in  series  and  a  combination  of 
resonant  circuits  connected  in  parallel. 

Figure  2  illustrates  the  four  possible  reactance  functions  which  are  distinguished  by 
their  behavior  at  zero  and  innnite  frequency.  The  series-type  networks  are  specified  in  the 
left-hand  column,  and  the  parallel  type  in  the  right-hand.  The  number  of  elements  in  each 
configuration  is  the  minimum,  and  equal  to  the  number  of  critical  frequencies  plus  1. 

The  operations  required  to  evaluate  an  expression  such  as 


are  to  be  performed  in  the  following  sequence:  (1)  multiply  —  by  —  •= 

f  ^         o)2jfc    — 

common  factor  (<w2Jb2  —  co2)  ;  (3)  replace  co  by  cx^jt. 


;  (2)  cancel  the 


6-36  PASSIVE   CIRCUIT  ELEMENTS 

Illustration. 


jw(o)22  —  W2)(co42  —  (02) 
This  is  realizable  with  the  second  pair  of  networks.    For  the  series  type: 


For  the  parallel  type: 


18.  PROPERTIES  OF  THE  IMAGE  PARAMETERS 

GENERAL  CONDITIONS.  In  the  pass  band  of  a  filter,  the  image  impedances  are 
positive  real  quantities,  or  resistances;  the  image  transfer  constant  is  pure  imaginary,  which 
signifies  phase  shift  with  zero  attenuation.  In  the  attenuating  region,  the  image  imped- 
ances are  pure  imaginary,  or  reactances;  the  transfer  constant  has  a  positive  real  compo- 
nent, which  signifies  attenuation. 

COINCIDENCE  CONDITIONS.  According  to  eqs.  (1)  and  (3),  the  image  impedance 
at  one  end  of  the  network,  and  the  transfer  function,  depend  only  on  the  short-  and  open- 
circuit  impedances  measured  at  that  end.  Typical  expressions  for  these  impedances  are: 

H^aJ  -  co2)  (q32  -  co2)  •  •  •  (am*  -  co2) 
41 


where  ai,  as,  05,  •  •  •  are  values  of  the  angular  frequency  at  which  the  short-circuited  net- 
work is  resonant,  and  a2,  a4,  •  •  •  are  values  of  the  angular  frequency  at  which  it  is  anti- 
resonant.  Similarly,  the  6's  with  odd  subscripts  are  the  resonant,  and  the  6'a  with  even 
subscripts  the  antiresonant,  angular  frequencies  of  the  network  when  the  far  end  is  open- 
circuited. 

The  expressions  for  the  image  impedance  and  transfer  constant  obtained  with  these 
impedance  functions  are: 

(ai2  -  co2)  •  •  •  (am*  -  co2)          (bi2  -  co2)  •  •  •  (&fe2  -  co2) 


-co2       (a22  -  co2)  •  •  •  (am_i2  -  co2)      (k2  -  co2)  -  -  -  (bk^  -  co2) 

Ei       (ai2  -  co2)  .-.  (aTO2  -  co2)        (bg*  -  co2)  •  •  »  (bk^  -  co2) 

Ha  '  (a22  -  co2)  ."  (am-i2  -  ^}  '    (6x2  -  co2)  •  -  •  (fefc2  -  co2)  U; 

The  statement  of  the  general  conditions  on  the  image  parameters  indicates  that  the 
network  transmits  freely  when  Zi*  is  positive  and  tanh2  0  is  negative;  it  attenuates  those 
frequencies  for  which  Zj*  is  negative  and  tanh2  0  is  positive.  In  general,  the  expressions 
(6)  and  (7)  change  sign  as  the  frequency  passes  through  each  a  and  6,  and  the  network  has 
a  multitude  of  pass  and  attenuating  bands.  In  order  that  the  network  be  a  filter  which 


PROPERTIES  OF  THE   IMAGE   PARAMETERS 


6-37 


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6-38  PASSIVE   CIRCUIT  ELEMENTS 

passes  one  continuous  band  of  frequencies  and  attenuates  all  other  bands,  it  is  necessary 
to  place  certain  evident  restrictions  on  the  critical  frequencies  of  its  short-  and  open-circuit 
impedances.  If,  for  example,  ai  =  61  =  coi,  the  factors  («i2  —  co2)  and  (&i2  —  co2)  cancel 
each  other  in  tanh2  6  but  form  the  factor  (coi2  —  co2)2  in  Zjf.  In  either  event,  as  co  passes 
through  the  value  coi  the  sign  of  the  expression  does  not  change.  It  follows  that  the 
elements  of  the  network  must  be  so  chosen  that  all  the  a's  are  equal  to,  or  coincide  with, 
6's  except  possibly  in  two  cases.  These  exceptional  cases  correspond  to  cutoff  frequencies 
at  which  the  network  changes  from  a  condition  of  free  transmission  to  attenuation,  or  vice 
versa.  These  transition  points  may  be  two  a's  or  two  b's  or  one  a  and  one  6. 

For  the  typical  short-  and  open-circuit  impedances  under  consideration,  it  is  seen  from 
(6)  and  (7)  that  Zi^  is  negative  and  tanh2  B  is  positive  at  zero  frequency.  That  is,  the 
filter  attenuates  zero  frequency.  This  condition  will  continue  over  a  frequency  band  pro- 
vided that  ai  =  61,  «2  —  bz,  or,  in  general,  a3  =  bj.  The  attenuation  band  is  terminated 
at  a  cutoff  frequency  by  locating  an  a  at  that  frequency  without  a  corresponding  6,  or 
vice  versa.  In  either  case,  if  a  continuous  pass  band  is  to  extend  beyond  the  cutoff,  it  is 
necessary  that  the  subsequent  coincidences  of  critical  frequencies  be  of  a  type  specified 
by  the  formula  a,  =  63±i,  since  the  sequence  either  of  a's  or  of  b's  has  lost  a  step  at  the 
cutoff.  The  pass  band  may  continue  to  infinite  frequency  or  it  may  be  terminated  at  a 
second  cutoff.  This  cutoff  may  be  specified  by  either  an  a  or  a  b.  Then  the  coincidences 
in  the  attenuating  band  above  this  cutoff  are  given  by  the  formula  a?  =  bj  or  a/  =  bj±z. 
The  first  relation  holds  if  one  cutoff  is  an  a  and  the  other  a  &;  the  second,  if  both  cutoffs 
are  a's  or  b's. 

Inspection  of  eqs.  (6)  and  (7)  leads  to  the  following  conclusions:  (a)  coincidences  of  the 
type  a,-  =  bj  and  a,  =  bj±%  produce  double  zeros  or  poles  in  Z/x2  but  cancel  out  in  tanh2  6; 
(6)  coincidences  of  the  type  a,  =  63-±i  produce  double  zeros  or  poles  in  tanh2  6  but  cancel 
out  in  Zi*\  (c)  critical  frequency  coincidences  of  the  first  type  produce  image-impedance- 
controlling  factors;  those  of  the  second  type  produce  transfer-constant-controlling  factors. 

SUMMARY  OF  PROPERTIES  OF  THE  IMAGE  PARAMETERS. 

1.  Tanh2  0  is  the  ratio  of  two  impedance  functions  of  a  reactive  network,  and  Z/2  is  the 
product  of  two  such  functions.    Tanh2  B  and  Zp  contain  only  double  zeros  and  poles  except 
possibly  for  two  zeros  or  poles,  which  are  simple. 

2.  The  simple  zeros  or  poles  represent  cutoff  frequencies  and  occur  at  positive  values 
of  co2.    They  are  the  same  for  the  two  expressions  except  for  the  possibility  that  either  or 
both  may  be  a  zero  in  one  expression  and  a  pole  in  the  other. 

3.  The  zeros  and  poles  of  tanh2  B  alternate  with  each  other  in  each  continuous  pass 
band;  the  zeros  and  poles  of  Zp  alternate  with  each  other  in  each  continuous  attenuating 
band.    The  step  between  bands  may  interrupt  the  alternation  since  the  cutoffs  may  both 
be  zeros  or  both  poles. 

Illustration.  Examples  of  expressions  for  the  short-  and  open-circuit  impedances  and  the  image 
parameters  are  given  in  Fig.  3.  The  frequency  pattern  is  a  convenient  schematic  representation  of 
these  functions.  It  is  a  plot  of  the  location  of  the  zeros  and  poles  of  a  function  in  the  frequency  scale. 
The  conventions  are:  circles  denote  zeros  of  the  function  (or  resonances  of  the  impedance);  crosses 
denote  poles  of  the  function  (or  antiresonances  of  the  impedance);  squares  represent  cutoffs.  These 
diagrams  serve  to  illustrate  the  properties  of  the  image  parameters  summarized  above.  The  network 
configurations  shown  in  the  table  are  "possible"  in  the  sense  that  they  will  be  obtained  for  an  appropriate 
choice  of  the  multipliers  HI  and  Hz  and  the  critical  frequencies  of  the  short-  and  open-circuit  impedances. 

19.  OPEN-CIRCUIT  TRANSFER  IMPEDANCE 

Two  important  filter  theorems,  due  to  H.  W.  Bode,  can  be  derived  from  the  properties 
of  the  open-circuit  transfer  impedance.  This  impedance,  designated  #012,  is  the  ratio  of 
the  voltage  appearing  across  the  open  output  terminals  of  the  network  to  the  current  fed 
into  the  input  terminals.  For  a  purely  reactive  network,  this  impedance  function  has  real 
coefficients  and  is  imaginary  at  real  values  of  frequency.  It  is  expressed  in  terms  of  the 
short-  and  open-circuit  impedances  by: 

By  use  of  eqs.  (l)-(3),  this  may  be  written: 

/i  \ 

(9) 


Since  #012  is  imaginary  along  the  real  frequency  axis,  Zoi22  must  be  negative  there.    Conse- 
quently, if  ZiZi^  changes  sign,  then  - — r-^-  —  1  must  change  sign  at  the  same  time,  and 

tann  \j 

conversely. 


IMAGE  IMPEDANCE  THEOREM  6-39 

In  the  pass  band,  the  image  impedances  are  resistances,  and  the  transfer  constant  is 
imaginary,  and  so  the  requirement  on  Z0i22  is  satisfied.  In  the  attenuating  band  where 
the  image  impedances  are  reactances,  and  tanh  6  is  real,  there  are  two  possibilities: 

(a)  If  the  image  impedances  are  reactances  of  the  same  sign,  the  product  Zi1Zr2  is 
negative.     Therefore,  tanh  6  must  be  equal  to  or  less  than  unity.    This  interval  will  be 
called  a  type  I  attenuating  band  and  will  be  designated  ABI. 

(b)  If  the  image  impedances  are  reactances  of  opposite  sign,  Z^Zj*  is  positive.    There- 
fore, tanh  6  must  be  equal  to  or  greater  than  unity.    This  interval  will  be  called  a  type  II 
attenuating  band  and  will  be  designated  ABII. 

20.  TRANSFER  CONSTANT  THEOREM 

The  transfer  constant  of  any  physically  realizable  filter  is  uniquely  determined  by  the  cutoff 
frequencies  and  by  the  frequencies  of  infinite  attenuation. 

The  frequencies  at  which  the  attenuation  is  infinite  are  the  roots  of  the  equation 
tanh  0—1.  In  practice,  they  are  found  by  determining  the  roots  of  the  equation 
tanh2  0—1  =  0.  The  number  of  roots  in  terms  of  c^2  is  equal  to  the  number  of  pole-zero 
intervals  of  tanh  6  in  the  pass  band  or,  equivalently,  the  total  phase  shift  in  radians  divided 
by  7T/2.  From  a  consideration  of  the  properties  of  the  open-circuit  transfer  impedance, 
it  is  deduced  that  the  admissible  roots  of  a  physically  realizable  filter  are  the  following: 
(a)  roots  of  even  multiplicity  located  at  positive  values  of  o^;  (6)  roots  of  even  multiplicity 
located  at  negative  values  of  oj2;  (c)  roots  of  even  multiplicity  located  at  conjugate  complex 
values  of  co2;  (d)  roots  of  odd  multiplicity  located  at  positive  values  of  co2.  As  the  later 
discussion  shows,  filter  sections  having  roots  of  even  multiplicity  are  the  rule  rather  than 
the  exception.  They  are  symmetrical  sections,  for  which  the  restrictions  on  the  image 
impedance  are  the  minimum.  They  are  usually  designed  to  have  the  simplest  image 
impedances. 

There  are  two  additional  restrictions  on  the  expression  for  tanh  6:  (a)  If  zero  or  infinite 
frequency  lies  in  a  pass  band,  tanh  &  must  have  a  zero  at  this  frequency;  (6)  if  zero  or 
infinite  frequency  lies  in  an  attenuating  band,  tanh  6  must  be  equal  to  or  less  than  unity 
at  this  frequency;  in  particular,  tanh  9  —  1,  if  this  frequency  lies  in  an  ABIT. 

The  significance  of  the  above  theorem  is  that  if  an  expression  for  tanh  6  is  found  which 
contains  the  chosen  cutoff  frequencies,  and  is  equal  to  unity  at  the  chosen  (admissible) 
frequencies  of  infinite  attenuation,  it  is  the  only  such  expression  that  does  exist,  and  it  will 
lead  to  a  physically  realizable  filter.  Such  an  expression  can  always  be  found. 

Illustration,    (a)  Test  the  expression 

tanh  6  • 

for  physical  readability.    The  equation  tanh  0—1  has  two  roots.    By  forming 

tanh2  0  -  1  =  ,     -1^»  -  0 

(1   —  6J2)2 

both  are  found  to  be  located  at  <w2  =  «.    Therefore,  this  expression  has  a  physical  representation, 
(b)  Test  the  expression 


for  physical  readability.    From 


it  is  seen  that  there  is  a  single  root  at  to2  =  —  1  and  another  at  o>2  =  «.    Therefore,  this  expression 
for  the  transfer  constant  is  not  realizable. 

21.  IMAGE  IMPEDANCE  THEOREM 

The  second  image  impedance  of  a  fitter  is  uniquely  determined  at  aU  frequencies,  except  for 
an  arbitrary  constant  multiplier,  by  the  transfer  constant  and  the  first  image  impedance. 

This  theorem  may  be  demonstrated  by  an  examination  of  the  properties  of  the  open- 
circuit  transfer  impedance.  The  formation  of  the  expression  for  the  second  image  imped- 
ance is  accomplished  by  the  application  of  the  following  rules: 

(a)  At  the  boundary  of  a  pass  band  and  ABI,  the  cutoff  factor  appears  in  the  numerator 
of  tanh  6  and  in  the  numerator  or  denominator  of  both  Z;l  and  Zj^. 


6-40 


PASSIVE   CIRCUIT  ELEMENTS 


(6)  At  the  boundary  of  a  pass  band  and  ABII,  the  cutoff  factor  appears  in  the  denom- 
inator of  tanh  6  and  in  the  numerator  of  one  image  impedance  and  in  the  denominator  of 
the  other  image  impedance. 

(c)  At  the  boundary,  ABI/ABII,  of  the  two  types  of  attenuating  bands,  the  equation 
tanh2  0  =  1  has  a  root  of  odd  multiplicity     One  image  impedance  must  have  a  zero,  or 
pole,  at  this  frequency  which  does  not  appear  in  the  other  image  impedance. 

(d)  All  other  zeros  and  poles  of  one  image  impedance  give  rise  to  corresponding  zeros 
and  poles  (or  poles  and  zeros)  in  the  second  image  impedance.    Attention  must  be  given 
to  the  requirement  that  the  zeros  and  poles  of  each  impedance  alternate  in  an  attenuation 
band. 

(e)  Roots  of  even  multiplicity  can  be  introduced  in  (tanh2  0  —  1)  without  regard  to  the 
image  impedances. 

Illustration.  The  expressions  for  the  transfer  constant  and  the  image  impedance  at  one  end  of  a 
band-pass  filter  are: 

tanh  Q  • 


h  -  ay* 


where  <ui  is  the  lower  cutoff,  and  &>2  the  upper  cutoff.  The  region  from  zero  frequency  to  o>i  is  an  .451, 
and,  by  rule  (a),  the  cutoff  factor  corresponding  to  «i  appears  in  the  denominator  of  Z/2.  The  region 
from  o>2  to  infinity  is  an  AJBII,  and,  by  rule  (6),  the  cutoff  factor  corresponding  to  W2  appears  in  the 
denominator  of  Zj^.  The  expression: 


has  one  root  in  w2,  and  that  is  at  infinity.    Thus,  there  are  no  other  internal  zeros  or  poles  in  Ziv 
Finally,  with  the  aid  of  rule  (d), 

»„-. 


The  elements  of  the  filter  may  be  found  from  the  short-  and  open-circuit  impedances: 


H 
1  o — IftRP- 


Zai  =  Zil  tanh  6  =>  Hju 


I'O- 


J.  1 


-o2' 


Kju 


FIG.  4.    Illustration  for  the  Method  of  Obtain- 

ing  the  Second  Image   Impedance  from  the 

Transfer  Function  and  the  First  Image  Im- 

pedance 


tanh  e 


_,  ,,,.-,,         ^  .       ^ 

Tbe  network  obtained  by  setting  K  = 

shown  in  Fig.  4. 


22.  THE  GENERAL  COMPOSITE  FILTER 

As  exemplified  by  the  preceding  illustration,  a  possible  filter  design  method  consists  of 
setting  up  physically  realizable  expressions  for  the  desired  transfer  constant  and  image 
impedance  at  one  end  of  the  filter.  Then,  the  expression  for  the  second  image  impedance 
is  found  with  the  aid  of  the  rules  associated  with  the  image  impedance  theorem.  The 
corresponding  short-  and  open-circuit  impedances  are  computed.  From  these,  it  is  possible 
to  find  the  elements  of  the  filter.  In  general,  this  may  require  some  ingenuity. 

The  design  method  which  is  used  most  frequently  is  based  on  the  fact  that  every  filter 
can   be   regarded   as   the 
combination  of  certain  ele-     1  °~ 
mentary   sections.      As         — 
shown  in  Fig.  5,  the  com- 
posite  filter   is   made  up 
of -TV  elementary  sections 
connected  in  tandem.  Sec- 
tion  A   provides   one   or 


lo- 


N 


-o2 


FIG.  5.    The  General  Composite  Filter 


more  frequencies  of  infinite  attenuation  (in  terms  of  to2)  and  has  the  image  impedance  Z^ 
specified  for  the  input  side  of  the  filter.  The  secondimage  im  pedance  Zia  is  deter- 
mined by  Zi1  and  the  roots  of  (tanh  6  =  1)  provided  by  section  A.  Section  B  has 


SYMMETRICAL  SECTIONS 


6-41 


the  image  impedance  Zja  to  match  A  and  provides  one  or  more  of  the  remaining  fre- 
quencies of  infinite  attenuation.  Its  other  image  impedance  is  Ziy  The  last  section  has 
the  image  impedance  Z/2,  consistent  with  the  specification  of  Z^  and  all  the  frequencies 
of  infinite  attenuation,  that  is,  all  the  roots  of  tanh  0=1. 

The  first  step  in  the  application  of  this  design  method  is  the  selection  of  the  terminating 
sections  to  furnish  the  desired  image  impedance.  They  provide  a  simple  image  impedance 
(i.e.,  one  without  impedance  controlling  factors)  at  their  inner  pairs  of  terminals.  These 
sections  correspond  to  A  and  N  in  Fig.  5.  The  transfer  constant  contributed  by  these 
sections  is  subtracted  from  the  required  overall  transfer  constant.  The  balance  is  supplied 
by  elementary  sections  of  known  transfer  constant  characteristics  which  have  simple  image 
impedances.  These  are  inserted  between  the  terminating  sections. 

An  alternative  design  procedure  is  applicable  where  it  is  convenient  to  establish  a 
physically  realizable  expression  for  the  transfer  function,  tanh  6,  which  meets  the  require- 
ments of  the  design  objective.  Such  a  case  arises,  for  example,  if  the  objective  is  to  attain 
linearity  of  phase  over  the  pass  band;  another  objective  which  can  be  handled  in  this  way, 
as  will  be  described  later  in  this  article,  is  to  provide  a  prescribed  minimum  of  attenuation 
over  a  specified  interval  of  the  attenuating  band.  The  next  step  in  the  design  is  the 
determination  of  the  frequencies  of  infinite  attenuation  by  solving  the  equation  tanh  6  —  1. 
Then  these  frequencies  are  assigned  to  the  appropriate  elementary  sections,  and  the  sections 
are  assembled  to  form  a  composite  filter. 

These  design  procedures  are  facilitated  if  there  is  available  a  list  of  the  characteristics 
and  element  values  of  the  elementary  sections  and  the  terminating  sections.  Such  a 
tabulation  should  contain  sections  which  provide  double  positive  and  negative  roots  in  co2, 
double  pairs  of  conjugate  complex  roots,  simple  positive  roots.  In  general,  the  double 
roots  correspond  to  the  elementary  symmetrical  sections  which  form  the  main  body  of 
.the  filter,  and  the  simple  roots  correspond  to  the  unsymmetrical  sections  which  are  used 
for  terminations.  The  tabulations  of  these  sections  are  discussed  in  the  following  two 
paragraphs. 

23.  SYMMETRICAL  SECTIONS 

The  problem  associated  with  the  design  of  symmetrical  filters  is  much  simpler  than  the 
general  design  problem.  The  configuration  which  is  convenient  for  analysis  is  the  lattice, 
shown  in  Fig.  6.  From  the  fundamental  eqs.  (l)-(3) 
for  the  image  parameters,  it  may  be  shown  that: 


(11) 

By  identifying  Zx  with  Zs\,  and  Z>y  with  ZQI,  the  anal- 
ysis in  terms  of  critical  frequencies,  given  in  article  18, 
is  directly  applicable.  The  restrictions  on  Zi  and  tanh 
0/2  are  simply  those  summarized  there  for  Zi  and 
tanh  0.  The  only  distinction  is  that  the  resulting 
transfer  constant,  0,  of  the  lattice  is  twice  the  value 

0/2  appearing  in  eq.  (11).  The  restrictions  contained  pia  6  The  General  Symmetrical 
in  the  transfer  constant  and  image  impedance  theo-  Lattice  Configuration 

rems  are  satisfied  since  all  the  roots  of  (tanh  0=1) 

are  of  even  multiplicity.  This  means  that  the  expressions  for  the  transfer  constant  and 
image  impedance  can  be  chosen  independently  except  for  the  cutoff  frequencies  which  are 
the  same  in  the  two  expressions.  The  branch  impedances  are: 


Zx  =  Zi  tanh  - 
2 


tanh  (0/2) 


(12) 


(13) 


These  impedances  satisfy  the  requirements  of  Foster's  theorem  and  may  be  developed  into 
the  structures  listed  in  Fig.  2. 

It  is  evident  that  these  branches  will  contain  a  large  number  of  elements  for  all  but  the 
simplest  filters.  Since  attenuation  is  obtained  by  bridge  balance,  these  elements  must 
be  held  to  close  limits  if  the  required  attenuation  is  great.  Consequently,  though  the 
lattice  is  much  used  in  theoretical  work,  it  is  usually  converted,  when  possible,  to  other 
configurations.  The  first  step  in  the  conversion  process  is  to  apply  the  concept  of  the 


6-42 


PASSIVE   CIRCUIT  ELEMENTS 


-ww- 


is 

2 


<  z 

FIG.  7.    Conversions  of  the  Symmetrical  Lattice 


SYMMETRICAL  SECTIONS 


6-43 


(1) 


(2) 


(3) 


(4) 


HH 

= 

c2 


mR 


mR 


mR 


mR 


(1  -  m2) 


2mvfcR 


Ci 


<    1 


V 


-& 

0  <m  <  oo 


'  /oo 

0  <m  <  I 


v 


- 


0  <  m  <  oo 


impedance 


fc 


Same  as  (1) 


V/c2  - 


Same  as  (3) 


jmf 


Same  as  (1; 


Same  as  (1) 


Same  as  (1) 


Special  case 
m  —  I 

/co-    «> 


1/2  =  short  circuit 


L2 


Ci  =  open  circmt 


=  C2 


FIG.  8.    Design  Information  for  Elementary  Symmetrical  Low-pass  Filter  Sections 


(D 


(2) 


(3) 


(4) 


II— « 


mR 


Ci 


Im-xfcR 


. 

/c2 
0  <  m  <  oo 


/c2 

<  oo 


Image 

impedance 

Zi 


Same  as  (I) 


Same  as  (3) 


tanh- 


w/c 


Same  as  (I) 


Same  as  (1) 


Special  case 
m  =  \ 


=  short  circuit 


Ci  =  C2 


LI  =  open  circuit 


LI  =  L2 


FIG.  9.    Design  Information  for  Elementary  Symmetrical  High-pass  Filter  Sections 


6-44 


PASSIVE   CIRCUIT  ELEMENTS 


*•••*          Q 


<u       ^ 

a   v 


^_^       o 

^   v 


LaaiJ 


o 

*s 


•is 


1 


SYMMETRICAL  SECTIONS 

I        ! 


s 


£v 


6-45 


-fh 


Hh 


^000  . 


f 


S^ 


sv 


4 


i 


lv 

020 


v 


.§*? 


•3  a  a 


5 
g 


2 
£ 


6-46 


PASSIVE  CIRCUIT  ELEMENTS 


^s 


€V 


e: 


CT=|=  N  g 

^SWU 1|— 


1      V 

vi 


-IS 


as 


SYMMETRICAL  SECTIONS 


6-47 


I  v 

e>     § 
S    V 


!* 

£  vi 


S  vi 


6-48 


PASSIVE   CIRCUIT  ELEMENTS 


composite  filter.  Then,  after  the  original  lattice  is  separated  into  a  combination  of  the 
simplest  lattices,  these,  in  turn,  are  converted  to  other  configurations.  Some  of  the  most 
useful  conversions,  based  on  Bartlett's  bisection  theorem,  are  given  in  Fig.  7.  (Broken 
lines  are  used  to  simplify  the  drawings.  It  is  understood  that  the  lattice  branches  repre- 
sented by  these  lines  are  duplicates  of  the  corresponding  ones  shown  explicitly.) 

ELEMENTARY  STRUCTURES.  The  elementary  symmetrical  sections  which  pro- 
vide the  double  positive  and  negative  roots  required  in  forming  a  composite  filter  are 
listed  in  Figs.  8,  9,  10.  Since  a  double  root  of  (tanh2  0—1)  corresponds  to  a  simple  root 

of  (  tanh2  -  —  1  J  ,  it  is  possible  to  use  the  more  convenient  expression,  tanh  (0/2) ,  in 

describing  these  sections.  Each  of  the  configurations  provides  one  double  peak  of  infinite 
attenuation.  The  associated  image  impedances  are  the  simplest  possible.  The  transfer 
constant,  0  —  a.  -f-  y/3,  where  a  is  the  attenuation  constant  in  nepers,  and  /3  is  the  phase 
constant  in  radians,  is  computed  from  the  expression  for  tanh  (0/2).  If  the  arithmetical 
value  of  this  expression  is  denoted  by  Q,  then,  in  the  pass  band: 

(9 
tanh  -  =  jQ 


In  the  attenuating  band: 


j3  =  2arctanQ;     (—> 


tanh-  =  £;     (Q  >  0) 


There  are  two  possibilities,  depending  on  the  value  of  Q  relative  to  unity.    Either: 

a.  =  2  arg  tanh  Q;     (Q  <  1) 
B  -  0 


o:  «s  2  arg  tanh  -  ;     (Q  >  1) 

|8    =     ±7T 

The  element  values  specified  in  these  tabulations,  as  well  as  all  that  follow,  apply  to  the 
filter  sections  as  drawn.    That  is,  each  section  is  considered  to  be  a  building  block  in  the 

composite  filter.  For  example,  if  two  lad- 
der-type mid-series  terminated  sections 
having  the  same  image  impedance  are 
joined  together,  the  intermediate  series 
impedance  becomes  equal  to  the  sum  of 
the  values  given  in  the  figures;  similarly, 
if  two  ladder-type  mid-shunt  terminated 
sections  are  joined  together,  the  inter- 
mediate shunt  admittance  becomes  equal 
to  the  sum  of  the  values  given. 

Figures  8  and  9  contain  the  design  in- 
formation for  low-pass  and  high-pass  sec- 
tions, respectively.  The  cutoff  frequency 
is  denoted  by  fe  and  the  frequency  of  in- 
finite attenuation  by  fw.  The  image  im- 
pedance is  equal  to  R  at  zero  frequency 
for  the  low  pass,  and  at  infinite  frequency 
for  the  high  pass.  The  sections  numbered 
(1)  and  (3)  provide  double  peaks  of  atten- 
uation at  real  frequencies.  They  are  the 
m-derived  sections  introduced  by  O.  J. 
Zobel.  The  special  case  for  which  m  =  1 
is  the  constant-jK"  section.  Sections  (2) 
and  (4)  provide  double  peaks  at  real  fre- 
quencies for  values  of  the  parameter  m 
lying  in  the  range  0  <  m  <  1,  and  dou- 
ble peaks  at  imaginary  values  of  frequency 
FIG.  11.  Combination  of  Two  Lattice  Sections  ^'e"  negative  values  of  co2)  for  values  of 
Which  Have  the  Same  Image  Impedance  m  greater  than  unity. 


SYMMETRICAL  SECTIONS 


6-49 


The  elementary  band-pass  sections  are  shown  in  Figs.  IQa  and  b.  The  lower  cutoff 
frequency  is  denoted  by  /i,  the  upper  cutoff  by  /2,  and  the  peak  of  infinite  attenuation 
by  /co-  The  image  impedance  is  equal  to  R  at  the  mid-band  frequency,  fm  =  Vf&.  A 


Type 


Image  Impedance 


Configuration 


Low-pass 


Rfc 


High-pass 


jfR 


-  A) 


jf(h-fi)R 


Band-pass 


FIG.  12.    Elementary  Constituents  of  the  General  Composite  Filter  Which  Provide  Attenuation  Peaks 
at  Complex  Values  of  Frequency 

uniform  definition  for  the  parameter  ra  is  used  throughout,  with  the  result  that  the  range 
of  values  is  extended  beyond  the  conventional  zero  to  unity.  The  odd-numbered  sections 
in  Fig.  10a  correspond  to  the  usual  m-derived  sections  which  have  a  peak  of  attenuation 
below  the  lower  cutoff.  The  special  cases,  for  which  m  =  1,  are  the  so-called  three- 
element  type  band-pass  sections  having  a  peak  at  zero  frequency.  The  lattice  sections 


6-50  PASSIVE   CIRCUIT  ELEMENTS 

provide  a  peak  below  the  lower  cutoff  for  0  <  m  <  1,  and  a  peak  above  the  upper  cutoff 
for  /2//i  <  m  <  oo .  For  the  range  1  <  m  <  /2//i,  the  peak  is  located  at  an  imaginary 
value  of  frequency.  In  all  cases,  the  phase  shift  is  ( —  x)  radians  at  f\  and  zero  at  /2. 

The  odd-numbered  sections  in  Fig.  106  correspond  to  the  usual  m-derived  sections  which 
have  a  peak  of  attenuation  above  the  upper  cutoff.  The  three-element  type  sections  having 
a  peak  at  infinite  frequency  are  special  cases,  for  which  m  —  fz/fi.  The  lattice  sections  are 
the  same  as  those  shown  in  Fig.  10a  except  that  the  branches  are  interchanged.  In  all 
cases,  the  phase  shift  is  zero  at  A,  and  (+w)  radians  at  /2. 

The  elementary  sections  which  provide  the  double  pairs  of  conjugate  complex  roots 
have  a  double  peak  of  attenuation  at  the  complex  value  of  frequency,  /„,  and  a  double 
peak  at  the  conjugate  value,  /,».  They  can  be  derived  by  combining  two  lattice  sections 
of  the  type  tabulated  in  Figs.  8—10.  The  method  for  combining  two  symmetrical  lattice 
sections  which  have  the  same  image  impedance  is  shown  in  Fig.  11.  By  using  the  defini- 
tions for  the  parameter  in  given  in  the  previous  figures,  a  complex  value,  m  =  mi  +  jmz, 
is  obtained  for  the  section  which  provides  the  peak  at  the  complex  value  of  frequency,  /^j 
the  conjugate  complex  value,  m  =  mi  —  jmz,  is  obtained  for  the  section  which  provides 
the  peak  at  the  conjugate  complex  value,  7^.  (The  real  part,  mi,  must  be  positive.)  The 
elements  of  the  individual  lattice  sections  are  proportional  to  m  and  m,  and  their  recipro- 
cals, and  therefore  are  complex  quantities.  However,  upon  combining  the  two  sections, 
the  resulting  elements  are  ordinary  coils  and  condensers.  These  sections  are  displayed 
in  Fig.  12. 

24.  UNSYMMETRICAL  SECTIONS 

The  unsymmetrical  sections  provide  the  simple  positive  roots  of  (tanh2  0  —  1)  in  terms 
of  to2.  Generally,  they  are  viewed  as  the  means  for  converting  the  simple  image  impedance 
of  the  main  part  of  the  composite  filter  into  an  image  impedance  which  approximates  as 
closely  as  required  to  a  constant  value,  equal  to  the  resistance  in  which  the  filter  is  termi- 
nated. For  low-  and  high-pass  filters,  the  conversion  is  from  a  constant-,??  type  "image 
impedance  to  one  having  one  or  more  impedance-controlling  frequencies.  For  band-pass 
filters,  there  is  a  greater  variety  of  possibilities.  However,  as  a  practical  matter,  the 
terminating  sections  are  usually  designed  to  convert  a  constant-.^  image  impedance  into 
one  having  one  or  more  pairs  of  impedance-controlling  frequencies.  The  product  of  the 
frequencies  making  up  a  pair  is  equal  to  the  square  of  the  mid-band  frequency.  For  com- 
pleteness, it  is  necessary  to  have  available  the  simple  sections  which  convert  a  three- 
element  type  image  impedance  into  the  geometrically  symmetrical  constant-^C  type. 

The  design  information  for  the  simple  sections  which  are  used  to  obtain  either  a  mid- 
series  or  mid-shunt  constant-.^  image  impedance  is  given  in  Fig.  13.  They  are  all  "half- 
sections,"  and  the  element  values  apply  to  the  sections  as  drawn.  Comparison  of  the 
formulas  for  the  element  values  and  image  impedance  of  the  three-element  band-pass 
sections  with  those  given  for  the  special  cases  in  Figs.  10a-Z>  shows  that  they  differ  by  a 
factor  which  is  a  function  of  the  cutoff  frequencies.  This  arises  from  the  fact  that  the 
symmetrical  sections  are  designed  to  give  an  image  impedance  equal  to  R  at  the  mid-band 
frequency,  while  those  in  Fig.  13  satisfy  this  condition  only  at  the  constant-^  end  of  the 
structure.  Hence,  the  impedance  level  of  one  or  the  other  must  be  changed  by  the  factor 
specified  in  the  lower  part  of  Fig.  13  before  they  can  be  joined  without  reflection.  In 
general,  the  level  of  the  symmetrical  sections  is  changed  so  that  the  constant-jfiT  impedance 
of  the  terminating  section  has  a  mid-band  value  equal  to  the  termination.  For  example, 
the  inductances  of  sections  7  and  8  of  Fig.  10a  are  multiplied  by  the  factor  C/2  +  /i)//2, 
and  the  capacitances  divided  by  this  factor,  if  the  section  is  joined  to  the  first  band-pass 
section  of  Fig.  13. 

m-DERIVED  SECTIONS.  The  design  information  for  terminating  sections  which 
present  a  constant-.^  image  impedance  at  one  end  and  an  image  impedance  having  one 
controlling  frequency  (or  one  geometrically  symmetrical  pair)  at  the  other  end  is  given 
in  Fig.  14.  It  follows  from  the  rules  associated  with  the  image  impedance  theorem  that 
the  low-  and  high-pass  sections  have  one  simple  attenuation  peak  at  the  impedance  con- 
trolling frequency,  and  the  band-pass  sections  have  a  geometrically  symmetrical  pair.  It 
is  convenient  to  use  a  universal  frequency  variable,  denoted  by  x,  in  the  description  of  all 
the  sections.  The  definition  of  this  variable  for  each  type  of  filter  is  included  in  the  figure. 
For  the  low-pass  sections,  the  pass  band  extends  from  x  —  0  to  +1,  and  the  attenuation 
band  from  x  —  -fl  to  plus  infinity.  For  the  high-pass  sections,  the  pass  band  extends 
from  2  =  0  to  2;=  —1,  and  the  attenuation  band  from  x  =  —1  to  minus  infinity.  For 
the  band-pass  sections,  the  pass  band  extends  from  x  —  —  lio  x  =  +1  with  the  mid-band 
at  x  =  0;  the  attenuation  band  above  the  upper  cutoff  extends  from  x  =  +1  to  plus 
infinity,  while  the  attenuation  band  below  the  lower  cutoff  extends  from  x  =  —  1  to  minus 
infinity. 


TJNSYMMETRICAL  SECTIONS 


6-51 


i 


ii 


^_L     UftJ 


+ 


+ 


e  i 


-i—4- 


Lo£Qj 


H- 


J I 


3 
5 


'T 


ii 


a  I 


^aS"-. 
-Ill6* 
a*".g 


6-52 


PASSIVE  CIRCUIT  ELEMENTS 


Mh_ 

ioooJ 


1 


—  \WL>  —  1|  — 


1 

O 


HI 


I 


p     « 

rf  co  f 
A  bO  c 

ill 


J< 

I 


•s 

eg 

I 
a 

1 


I 


I 


UNSYMMETRICAL  SECTIONS 


6-53 


The  choice  of  a  particular  value  of  the  parameter  m  is  dictated  by  the  image  impedance 
characteristic  that  is  desired  (on  the  assumption  that  the  location  of  the  associated  atten- 
uation peak  is  satisfactory).  Several  characteristics  are  shown  in  Fig.  15  including  the 
constant-/?  type,  for  which  m  =  1.0.  These  curves  exhibit  the  course  of  the  mid-series 
image  impedance  and  the  mid-shunt  image  admittance  as  a  function  of  the  frequency 
variable,  x.  A  generally  satisfactory  value  of  the  parameter  is  m  =  0.6.  It  is  seen  that 
the  image  impedance  of  this  section  remains  within  about  4  per  cent  of  a  constant  value 
over  87  per  cent  of  the  pass  band.  The  actual  impedance  measured  at  the  terminals  of 

1.2 
1.1 


1.0 
0.9 
0.8 
0.7 

I  1  0-6 

II  0.5 

IJL 

0.4 
0.3 
0.2 
0.1 


V 


~0        0.1        0.2       0.3       0.4       0.5       0.6       0.7       0.8       0.9       1.0 
Frequency  Variable  x 

FIG.  15.    Image  Impedance  Characteristics  of  Several  m-derived  Filter  Sections 

the  filter  is  identical  with  the  image  impedance  only  if  the  other  end  of  the  filter  is  termi- 
nated in  the  image  impedance  at  that  end.  A  general  formula  for  computing  this  im- 
pedance is 


where  Zj  is  the  image  impedance  at  the  end  of  the  filter  for  which  the  driving  point  im- 
pedance Z  is  being  calculated;  Z/2  is  the  image  impedance  at  the  far  end  which  is  termi- 
nated by  the  load  impedance  Rt',  and  6  is  the  total  image  transfer  constant  of  the  complete 
filter.  Evidently,  if  Z/2  =  Rt,  then  Z  =  Z/r 

TWO-FREQUENCY  CONTROL  SECTIONS.  Terminating  sections  which  present 
an  image  impedance  having  two  controlling  frequencies  may  be  derived  in  a  number  of 
ways  and  exist  in  various  forms.  The  sections  listed  in  Fig.  16  serve  as  a  direct  transition 
from  a  constant-X  image  impedance  to  a  "two-frequency  control"  image  impedance. 
They  are  units  consisting  of  four  branches  which  cannot  be  separated  at  any  internal 
junction,  even  though  they  have  the  structural  appearance  of  1  1/2  section  m-derived 
filters.  The  low-  and  high-pass  sections  have  two  simple  attenuation  peaks,  and  the 
band-pass  have  two  geometrically  symmetrical  pairs  of  simple  peaks.  The  behavior  of 


6-54 


PASSIVE    CIRCUIT  ELEMENTS 


i 


•ss 


III 


o^ 


UNSYMMETRICAL  SECTIONS 


6-55 


tf 

i"B 


g     g 


+ 


'5 


g 

A 


S 

§ 

1 


I 
> 


Ci 


JfS 
>SVVs 


V     8 


P 
S 


EH 


^ 

2 


Q 
«5 


6-56 


PASSIVE   CIBCUTT  ELEMENTS 


the  image  impedance  depends  on  the  choice  of  the  parameters,  mi  and  mz.  Since  the  usual 
objective  is  to  obtain  an  impedance  which  approximates  closely  to  a  constant  resistance, 
these  parameters  may  be  determined  most  easily  by  the  method  described  in  the  next 
section. 

25.  TCHEBYCHEFF  TYPE  CHARACTERISTICS 


In  the  preceding  two  sections,  attention  has  been  directed  at  the  building  blocks  that 
make  up  a  composite  filter.  These  elementary  units  can  be  assembled  to  give  a  great 
variety  of  characteristics.  The  characteristics  desired  in  a  filter  design  depend,  of  course, 
on  the  particular  use  for  which  the  filter  is  intended.  Many  applications  include  the  re- 
quirement that  the  image  impedance  should  be  substantially  constant  over  a  prescribed 
portion  of  the  pass  band.  Another  frequent  specification  is  that  the  attenuation  exceed  a 
given  value  at  all  frequencies  more  than  a  certain  distance  beyond  the  cutoff.  An  equiva- 
lent statement  of  these  requirements  is:  (1)  the  image  impedance  function,  Zj/Rt,  should 

approximate  to  unity  over  a  pre- 
assigned  interval  of  the  pass 
band;  (2)  the  image  transfer  func- 
tion, tanh  6,  should  approximate 
to  unity  over  a  preassigned  inter- 
val of  the  attenuating  band. 

A  desirable  approximation  is 
one  for  which  the  maximum  de- 
partures of  the  function  from 
unity  are  the  minimum.  This  is 
known  as  the  Tchebycheff:  ap- 
proximation, proposed  originally 
by  W.  Cauer  for  this  application. 
In  this  type  of  approximation, 
as  illustrated  in  Fig.  17,  the  func- 


Scale for  2T/R<  or  Rf/Z]. 


Scale  for  tanh  0/2  or  coth  0/2 


l»i 


-- 


-,  —  -7-  for  low-pass  filter 
Jc 

x  =  — 7  for  high-pass  filter 


- 

~  /l) 


r  band-pass  filter 


FIG.  17.     The  Tchebycheff  Type  of  Approximation  to  a  Con- 
stant Value 


tion  winds  about  unity  in  such  a 
way  that  all  the  maximum  values 
(U)  are  equal,  and  all  the  mini- 
mum values  (I/  (7)  are  also  equal. 
Unity  is  the  geometric  mean  of 
the  maximum  and  minimum  val- 
ues. The  number  of  intersec- 
tions with  unity  is  determined  by 
the  degree  of  the  function. 
ATTENUATION  CHARACTERISTIC.  The  image  transfer  function,  tanh  (6/2},  of  a 
symmetrical  filter  is  written  in  terms  of  the  frequency  variable,  \/x.  The  critical  frequen- 
cies and  the  constant  multiplier  are  so  chosen  that  the  function  is  constrained  to  lie  between 
the  limits  U  and  1/Z7,  in  the  frequency  range  from  l/k  to  infinity,  as  shown  in  Fig.  17. 
The  intersections  with  unity  correspond  to  peaks  of  infinite  attenuation;  the  equal  values 
of  minimum  attenuation,  AQ,  are  equal  either  to  2  arg  tanh  U,  or  2  arg  coth  U.  For  a 
given  number  of  filter  sections,  there  is  a  definite  relation  between  k  and  AQ. 

This  relation  is  presented  in  chart  form  in  Fig.  18.  The  parameter,  JV,  is  equal  to  the 
number  of  transfer  constant  controlling  factors  in  the  image  transfer  function.  The 
number  of  m-derived  low-  and  high-pass  filter  sections,  Figs.  8  and  9,  is  equal  to  (N  -f-  1)  ; 
the  number  of  m-derived  band-pass  sections,  Figs.  10a  and  106,  is  [2(N  -f-  1)].  The 
horizontal  scale  of  this  chart  is  spread  out  considerably  for  values  of  k  near  unity  by  using 
the  variable  a  defined  by  the  equation  k  =  sin  a.  The  data  required  for  making  up  this 
chart  may  be  obtained  from  the  following  set  of  computations: 


(1) 

(2) 
(3) 
(4) 


a  =  arc  sin  k 


1  - 


2e 


AQ  ±  20(N  +  1)  log-  -6.0 
Q. 


(in  db) 


This  relation,  derived  by  S.  Darlington,  gives  a  result  good  to  within  0.1  db  provided  AQ 
is  greater  than  about  6  db.  A  table  of  log  q  vs.  a.  is  contained  in  Funktionentafeln  by 
Jahnke  and  Emde.  This  may  be  used  in  place  of  (2)  and  (3) . 


TCHEBYCHEFF  TYPE  CHARACTERISTICS 


6-57 


It  is  not  likely  that  arbitrarily  chosen  values  of  AQ  and  a  will  lie  on  one  of  the  curves. 
Thus,  a  compromise  must  be  made  between  minimum  attenuation  and  the  interval  of 
the  attenuating  band  that  is  covered.  For  example,  if  a  filter  is  to  be  designed  to  attenuate 
frequencies  above  x  =  1.10  by  about  60  db,  a  choice  must  be  made  among  the  possibili- 
ties: A0  =  52  db,  a  =  65°,  I/A  =  1.10,  A"  =  2;  AQ  =  60  db,  a  =  58°,  l/k  =  1.18,  N  =  2; 


100 
90 

80; 


<60 


< 
E40 


20 


10 


\ 


N 


X'h 


K 


0.01 


0.05 


0.1 


0.2  -5 


20.0 


50°       55° 


60°       65°        70° 
a=arc  sin  k 


75°       80°       85" 


FIG.  18.    Design  Chart  for  Determining  the  Number,  N,  of  Frequency  Control  Factors  Required  for 
a  Specified  Reflection  Coefficient  or  Minimum  Attenuation 

AQ  —  60  db,  a  =  73°,  l/k  =  1.045,  N  =3.  It  is  customary  to  design  to  a  value  of  min- 
imum attenuation  which  is  6  db  greater  than  the  desired  minimum  insertion  loss.  This 
allows  for  3  db  reflection  gain  at  each  end  of  the  filter,  which  is  the  maximum  value  that 
may  be  realized. 

The  design  formulas  for  the  element  values  are  expressed  in  terms  of  the  impedance  level, 
R,  the  cutoff  frequency  or  frequencies,  and  the  parameter,  m.  At  this  point  it  is  assumed 
that  the  first  two  quantities  are  known.  The  values  of  m  are  determined  from  the  fre- 
quencies of  infinite  attenuation.  They  are  given  by: 


s  -  1,  2,  ••-,  A"+  1 


where  sn  denotes  an  elliptic  sine  function  of  modulus  k,  and  K  is  the  complete  elliptic 
integral  of  the  first  kind.  Each  choice  of  the  index  s  specifies  an  attenuation  peak  at  the 
frequency  corresponding  to  xs. 

The  evaluation  of  the  sn  function  can  be  performed  by  means  of  elliptic  integral  tables. 
However,  a  preferred  method  is  to  use  the  approximation: 


ur 


r  J 


where 


2u  •• 


1  -  VJfc 

1  +  Vk 

2s  -  1 

2(A^  -f  1) 


=  1,  2,  ••- 


6-58  PASSIVE   CIRCUIT  ELEMENTS 

The  actual  peak  frequencies  used  in  the  definition  of  m  in  Figs.  8,  9,  10  are  obtained 
from  the  normalized  frequency  for  each  type  of  niter  from  the  following  relations: 

(a)  low-pass  /w  =  fcxs 

f 
(6)   high-pass  fw  —  — 


(c)    band-pass  /«  -  fn[Dx,  +  Vl  +  (Da;,)2];     (>/2) 


/_  =  fm[-Dx,  +  Vl  +  (Dxsn     ( <A) 
D=f±^ 


The  band-pass  attenuation  characteristic  is  geometrically  symmetrical  about  the  mid- 
band  frequency.  That  is,  the  attenuation  at  a  frequency,  fx,  above  the  upper  cutoff  is 
the  same  as  the  attenuation  at  the  frequency  /_»  =  fm2/fx  below  the  lower  cutoff.  Instead 
of  designing  the  four-element  band-pass  sections,  Fig.  10,  it  is  more  convenient  and 
economical  to  use  the  six-element  sections  described  in  Fig.  14.  Two  half-sections  must 
be  joined  together  to  realize  the  performance  predicted  by  the  above  design  method.  It 
is  to  be  noted  that  the  parameter,  m,  is  expressed  directly  in  terms  of  xs. 

IMAGE  IMPEDANCE  CHARACTERISTIC.  The  image  impedance  function  is  written 
in  the  form  Zi/Rt  in  terms  of  the  frequency  variable  x.  The  terminating  resistance  is 
denoted  by  Rt-  The  critical  frequencies  and  the  constant  multiplier  R  are  chosen  in  such 
a  way  that  the  function  remains  within  the  limits  U  =  R  and  \/U  =  1/R  over  the  fre- 
quency range  from  zero  to  k,  as  shown  in  Fig.  17.  Perfect  match  points  correspond  to  the 
intersections  with  unity.  Because  of  the  reciprocal  nature  of  the  departures  from  unity, 
it  is  evident  that  the  maximum  departures  of  the  magnitude  of  the  reflection  coefficient 
from  zero  are  all  equal  over  the  approximation  interval. 

The  relation  between  the  reflection  coefficient,  p,  and  the  pass-band  interval,  &,  within 
which  the  reflection  coefficient  does  not  exceed  the  prescribed  limit  is  presented  in  Fig.  18. 
(The  scale  chosen  for  the  reflection  coefficient  is  not  a  convenient  one.  However,  it  can 
be  easily  transformed  into  the  db  scale  in  accordance  with  the  relation  AQ  =  20  log 
(100/p),  where  p  is  in  per  cent,  and  AQ  in  decibels.)  The  parameter  N  is  equal  to  the 
number  of  impedance  controlling  factors  in  the  expression  for  the  image  impedance.  For 
the  constant-X  sections,  Fig.  13,  N  —  0;  for  the  usual  m-derived  sections,  Fig.  14,  N  =  1; 
and  for  the  "two-control"  terminating  sections  shown  in  Fig.  16,  N  =  2. 

The  element  values  for  the  terminating  sections  depend  on  the  cutoff  frequencies,  the 
impedance  level,  and  the  parameter  m.  It  is  assumed  here  that  the  cutoff  frequencies 
are  known.  The  impedance  level  is  specified  by: 

R_  =  1  +  p  1  -  p 

Rt  ~  1  -  P    °r    1  +  P 

where  Rt  is  the  terminating,  or  load,  resistance,  and  p  is  the  absolute  value  of  the  reflection 
coefficient  (not  in  per  cent).  The  first  relation  is  used,  for  a  mid-series  type  impedance 
when  N  is  even,  and  for  a  mid-shunt  type  when  N  is  odd;  the  second  relation  is  used  for 
the  other  two  possibilities. 

The  values  for  m  are  determined  in  terms  of  the  frequencies  of  infinite  attenuation. 
These  are  specified  by: 

i  /     +TT          \ 

t  =  1,  2,  --,N 


The  following  approximation  is  usually  more  satisfactory  for  evaluating  these  peak  fre- 
quencies than  the  use  of  a  table: 


xt      Vfcl  +  uv  +  u?~v 
where 

1  -  Vk 

2u  =  -  — 

1  +  Vfc 
and 


TCHEBYCHEFF  TYPE  CHAKACTEEISTICS 


6-59 


V 


1 


\ 


I 
E 


qp'uoijsnusuv 


2       °3 


qp  'u 


6-60 


PASSIVE   CIRCUIT  ELEMENTS 


T 


c  - 
'c    o 

P     0> 


\ 


II 


I— 

1 


g 

•qp  'u 


-  ID 

rH 
10 


o 

S 
6 


MAYER'S  THEOREM  6-61 

Illustration.  The  application  of  this  techniaue  to  the  design  of  a  band-pass  filter  serves  to  clarify 
some  of  the  details  of  the  procedure.  It  is  assumed  that  the  lower  cutoff  is  12  kc  and  the  upper  cutoff 
is  16  kc.  The  minimum  insertion  loss  should  be  at  least  52  db  above  16.4  kc,  and  below  11.6  kc.  The 
reflection  coefficient  should  be  less  than  1  per  cent  between  12.1  and  15.9  kc. 

The  terminating  sections  are  designed  first,  so  that  the  attenuation  which  they  contribute  may  be 
deducted  from  the  58  db  requirement  (allowing  6  db  for  reflection  gain).  From  the  curve,  X  —  2,  on 
Fig.  18,  it  is  seen  that  a  1  per  cent  reflection  coefficient  can  be  realized  over  the  portion  of  the  pass 
band  corresponding  to  a  —  76°  by  using  the  sections  shown  in  Fig.  16.  The  actual  frequency  range 
lies  between  the  limits  /+  =  15,931  and  /_=  12,050  corresponding  to  a;  =  k  =  sin  76°  =  0.9703,  as 
computed  from: 


The  plus  sign  gives  the  frequency  above  mid-band,  and  the  minus  sign  the  frequency  below  mid-band. 
The  quantity  D  is  the  relative  bandwidth  equal  to  Cfc  —  fi)/2fm.  The  distance  between  the  upper 
and  lower  frequencies  is  proportional  to  the  absolute  band  width,  i.e.,  (/+  —  /_)  —  (fz  ~~  /i)x. 

The  impedance  level  of  the  section,  for  p  =  0.01,  is  found  to  be  R  =  1.020.Kf.  For  precise  results,  a 
value  for  the  reflection  coefficient,  superior  to  that  read  from  Fig.  18,  may  be  calculated  from  the  equa- 
tion p  —  Sq  .  The  image  impedance  characteristic  is  similar  to  the  curve  displayed  in  Fig.  17  with 
U  =  1.020  and  k  —  0.9703.  The  negative  half  extending  from  zero  to  minus  one,  corresponding  to  the 
interval  from  mid-band  to  the  lower  cutoff,  is  a  duplicate  of  the  part  shown. 

The  final  step  in  the  design  is  the  evaluation  of  the  peak  frequencies  by  means  of  the  approximation 
given  above,  with  the  index  r  =  1/3  and  2/3.  The  results  are  xi  =  1.0351  and  x%  =  1.3484;  the  corre- 
sponding values  of  the  parameter  are  mi  =  0.25838  and  mz  =  0.67082.  The  element  values  are  ob- 
tained from  Fig.  16.  An  appropriate  configuration  is  shown  in  Fig.  19a,  as  well  as  the  attenuation 
characteristic  of  the  two  terminating  sections,  which  is  obtained  by  doubling  the  result  calculated 
from  the  formula  in  Fig.  16. 

That  part  of  the  complete  filter  which  is  between  the  terminating  sections  supplies  the  balance  of 
the  required  attenuation,  i.e.,  the  difference  between  58  db  and  that  shown  in  Fig.  19o.  This  unit  may 
be  designed  by  fitting  together  the  attenuation  characteristics  of  four  m-derived  sections  (Fig.  10)  ; 
several  trials  may  be  necessary.  It  is  interesting  to  see  how  the  design  process  under  discussion  leads 
directly  to  a  satisfactory  choice  for  these  intermediate  sections. 

The  chart,  Fig.  18,  is  entered  with  AQ  =  58  db.  This  attenuation  can  be  obtained  with  three  sections 
at  frequencies  above  and  below  those  corresponding  to  a  =  60°.  The  limits  are  found  to  be  f+  =  16,357 
and  /_  =  1  1  ,738.  The  values  for  the  peak  frequencies  are  computed  from  the  approximation  given  above, 
with  the  index  r  =  1/6,  3/6,  5/6-  They  are  x\  =  1.1742,  and  xi  =  1.4142,  and  xz  =  3.333;  the  corre- 
sponding values  of  the  parameter  are  mi  =  0.52415,  and  mi  —  0.70711,  and  7713  =  0.95393.  The 
attenuation  characteristics  of  the  three  sections  are  calculated  from  the  formula  given  in  Fig.  14  (this 
formula  applies  to  a  half  -section  so  that  the  result  must  be  doubled).  Figure  196  is  a  sketch  of  the 
complete  characteristic.  Two  of  the  possible  configurations  are  shown;  the  first  consists  of  sections 
chosen  from  Fig.  10,  and  the  second  is  taken  from  Fig.  14  (an  equivalent  shunt  branch  is  used). 

The  sum  of  the  characteristics,  Figs.  19a  and  &,  is  far  in  excess  of  the  attenuation  required.  It  is 
observed  that  the  terminating  section  has  a  peak  at  £2  —  1.35.  This  is  close  to  the  peak,  x»  =  1.41,  of 
one  of  the  attenuation  sections.  Consequently,  this  latter  section  may  be  omitted  from  the  intermediate 
part  of  the  complete  filter  without  affecting  greatly  the  minimum  attenuation  of  58  db.  There  is  a 
considerable  increase  in  attenuation  between  the  cutoffs  and  the  limit  frequencies,  ±1  /k,  owing  to  the 
first  peak  in  the  terminating  sections.  However,  a  large  part  of  this  surplus  will  be  lost  because  of 
dissipation  in  the  components.  The  overall  attenuation  is  plotted  on  an  arc  sin  frequency  scale  in  Fig.  19c. 


26.  MAYER'S  THEOREM 

The  effect  of  uniform  dissipation  in  the  elements  on  the  characteristics  of  a  reactive  net- 
work can  be  estimated  readily  by  means  of  Mayer's  theorem.    This  states  that: 


A  and  B  represent  the  real  and  imaginary  parts,  respectively,  of  a  network  characteristic 
in  the  absence  of  dissipation.  Ad  and  Bd  are  the  corresponding  quantities  when  dissipation 
is  present.  As  usual,  <a  is  2ir  times  the  frequency.  The  average  dissipation  is: 


Q      2\QL      Qc) 

where  Qz,  is  the  average  aL/r  for  the  coils,  and  Qc  is  the  average  eaC/g  for  the  condensers. 
Frequently,  the  condensers  are  considered  relatively  non-dissipative,  so  that  Q  =  2Q&. 

Since  these  relations  come  from  a  Taylor's  series  development  of  the  network  function, 
they  are  usable  only  in  the  regions  where  the  function  is  well  behaved.  For  example,  if 
A  and  B  represent  image  attenuation  and  image  phase  shift,  the  approximation  fails  in 


6-62  PASSIVE   CIRCUIT  ELEMENTS 

the  neighborhood  of  the  cutoffs  and  the  attenuation  peaks;  on  the  other  hand,  if  A  and  B 
represent  the  insertion  loss  and  insertion  phase  shift,  the  approximation  holds  everywhere 
except  near  the  loss  peaks;  a  similar  remark  obtains  for  driving  point  impedance,  current 
ratio,  voltage  ratio,  etc.,  functions. 

These  formulas  lead  to  some  useful  general  conclusions.  They  indicate  that,  to  a  first 
approximation,  the  change  produced  in  the  real  component  of  a  network  characteristic  by 
dissipation  is  proportional  to  the  slope  of  its  imaginary  component,  and  vice  versa.  It  is 
particularly  interesting  to  note  the  effect  in  respect  to  characteristics  which  approach  the 
ideal.  For  example,  the  approximation  to  an  ideal  filter  is  designed  to  have  a  constant 
(zero)  attenuation  and  a  phase  shift  which  varies  linearly  with  frequency  over  most  of  the 
pass  band.  It  is  to  be  expected  that  the  phase  characteristic  will  be  changed  very  little 
by  dissipation  since  the  slope  of  the  attenuation  is  zero.  On  the  other  hand,  the  attenua- 
tion characteristic  will  be  affected  when  dissipation  is  present.  However,  over  the  fre- 
quency range  for  which  Q  is  proportional  to  frequency,  the  attenuation  will  be  constant 
and  proportional  to  the  phase  slope.  Consequently,  the  filter  introduces  a  flat  loss  but 
does  not  introduce  distortion. 

REFERENCEvS 

Bartlett,  A.  C.,  The  Theory  of  Electrical  Artificial  Lines  and  Filters.    John  Wiley  (1930). 

Bode,  H.  W.,  A  General  Theory  of  Electric  Wave  Filters,  J.  Math,  and  Phys.t  Vol.  13,  275-362 

(November  1934). 

Bode,  H.  W.,  and  R.  L.  Dietzold,  Ideal  Wave  Filters,  B.S.T.J.,  Vol.  14,  215-252  (April  1935). 
Cauer,  W.,  Siebschaltungen,  V.D.I.  Verlag  G.m.b.H.,  Berlin  (1931). 
Cauer,  W.,  New  Theory  and  Design'of  Wave  Filters,  Physics,  Vol.  2,  242  (April  1932). 
Darlington,  S.,  Synthesis  of  Reactance  4-Poles,  /.  Math,  and  Phys.,  Vol.  18,  257-353  (September  1939). 
Foster,  R.  M.,  A  Reactance  Theorem,  B.S.T.J.,  Vol.  3,  259-267  (April  1924). 
Guillemin,  E.  A.,  Communication  Networks,  Vol.  II.    John  Wiley  (1935). 
Zobel,  O.  J.,  Theory  and  Design  of  Uniform  and  Composite  Electric  Wave  Filters,  B.S.T.J.,  Vol.  2, 

1-46  (January  1923). 

RADIO  ANTENNAS 

By  J.  C.  Schelleng  * 

GENERAL  FUNCTION  AND  DESCRIPTION.  The  antenna  is  the  means  of  coupling 
between  the  medium  of  propagation  and  the  transmitter  or  receiver.  Its  purpose  is  to 
convert  power  into  outgoing  electromagnetic  waves,  or  to  extract  power  from  incoming 
waves.  These  two  processes  are  reciprocal,  but  other  factors  such  as  discrimination  against 
static  introduce  new  requirements  which  differentiate  a  good  transmitting  from  a  good 
receiving  antenna. 

Not  only  should  an  antenna  be  an  efficient  converter  between  radiant  and  guided 
energy,  but  it  should  also  be  most  effective  in  the  direction  of  the  station  with  which  it  is 
communicating.  At  the  receiving  end,  directivity  performs  two  useful  functions.  The 
first  is  that  of  impressing  on  the  early  stages  of  the  receiver  a  relatively  intense  signal  so 
as  to  override  the  inherent  circuit  or  tube  noises.  The  second  and  often  the  more  important 
is  that  of  discriminating  against  electrical  disturbances  in  the  medium,  such  as  atmospheric 
noise,  thereby  increasing  the  ratio  between  signal  and  noise  components.  If  the  atmos- 
pherics should  arrive  from  the  direction  of  the  signal,  then,  of  course,  no  advantage  would 
result  from  this  means.  When,  on  the  average,  they  arrive  with  equal  intensity  from  all 
directions,  the  advantage  is  considerable.  If  it  happens,  as  in  communication  between 
Europe  and  America,  that  a  very  small  part  of  the  total  noise  arrives  from  the  direction 
of  the  desired  signal,  a  very  great  advantage  is  realized.  Indeed,  because  of  their  ability 
to  discriminate  against  atmospherics  arriving  from  the  rear,  it  is  sometimes  the  practice 
to  use  as  receiving  antennas  devices  that  would  ordinarily  be  relatively  poor  radiators. 

Practical  antennas  differ  widely.  Perhaps  the  simplest  is  a  straight  vertical  wire  as 
shown  in  Fig.  la,  6,  c,  and  d.  For  the  lowest  frequencies  used  in  radio  the  length  is  a  very 
small  fraction  of  a  wavelength,  but  with  higher  frequencies  the  vertical  wire  may  exceed  a 
half-wavelength.  The  dotted  lines  indicate  roughly  the  distribution  of  current.  Several 
antennas  may  be  arranged  to  form  a  directive  array  as  shown  in  Fig.  H.  An  antenna 
one-half  wave  long  (Fig.  Ig)  is  known  as  a  half-wave  linear  antenna. 

When  the  frequency  is  low  (long  waves) ,  the  antenna  may  take  the  form  of  an  L  or  a  T, 
a  hundred  or  so  feet  high  and  several  hundred  long  (see  Fig.  le  and  /).  The  horizontal 
portion  is  known  as  the  flat-top,  the  vertical  as  the  lead-in.  Frequently  a  large  coil  of  wire 
is  used  as  an  aerial  as  shown  in  Fig.  Ik.  It  is  known  as  a  loop. 

*  In  this  revision  a  considerable  amount  of  material  from  the  article  Radio  Antennas  (7-57)  appearing 
in  the  1936  Edition  of  the  Pender-Mcllwain  Handbook,  and  written  by  G.  C.  Southworth,  has  been  used. 


KADIO  ANTENNAS 


6-63 


In  the  highest  decade  of  the  radio  spectrum  (in  1949  the  practical  limit  seems  to  be  of 
the  order  of  30,000  megacycles  per  second,  wavelength  1  cm),  the  low-frequency  technique 
associated  with  radiation  "from  the  outside"  of  the  conductors  has  become  relatively 


Grounded  Antennas 


Ungrounded  Antenna 
Ungrounded  Antennas  Without  Loading  With  Loading  Loop  Antenna 

L    <y)/=5>\  m7=— 


Broadside  Acray 


0 


I") 


FIG.  1.    Representative  Forms  of  Antennas 


Sidp         ~*'2~' 

Elevation  Front  E!evat!on 

Y\\i  aim  IK, 
in  ji  i  i  IN  iHf 


difficult  owing  to  the  small  size  of  the  wavelength,  and  antenna  designers  tend  to  draw  their 
inspiration  from  wave-guide  principles  and  from  optics.  Typical  antennas  now  are  found 
to  employ  wave-guide  apertures  or  horns,  parabolic  reflectors  or  lenses,  as  well  as  many 


(a) 


FIG.  2.    Forms  of  Quasi-optical  Antenna 


of  the  older  techniques.  Figure  2a  illustrates  radiation  from  the  open  end  of  a  circular 
wave  guide,  2b  shows  a  pyramidal  horn  formed  by  tapering  from  a  small  rectangular  wave 
guide,  and  2c  shows  a  linear  half-wave  antenna  at  the  focus  of  a  paraboloidal  antenna. 


6-64  PASSIVE   CIRCUIT  ELEMENTS 

CLASSIFICATION.  Table  1  is  an  attempt  to  classify  the  many  diverse  forms  of 
antennas  which  have  found  use.  Ignoring  distinctions  between  transmitting  and  receiving, 
it  gives  one  type  of  classification  based  on  general  form  rather  than  specific  application. 
In  spite  of  a  degree  of  artificiality  it  covers  most  of  the  common  types.  The  relatively 
non-directional  appear  early  in  the  list;  the  highly  directional  are  found  at  the  end. 

Table  1.    Classification  of  Antennas 

Linear  Conductor  Antennas 

L,  T,  umbrella,  multiple-tuned  electric  dipoles,  "vertical  radiators,"  whip,  trailing- wire,  vertical 
wire,  horizontal  wire,  wave  antenna,  loops,  etc. 
Wave-guide  Antennas 

Small  wave-guide  apertures: 

Open-end  wave  guide. 

Slot  antennas. 

Various  reflector  or  lens  feeds,  etc. 
Large  wave-guide  apertures:  Horns. 
Dielectric  antennas:  Polyrods. 
Quasi-optical  Devices:  Antennas  Using  Reflectors  or  Lenses  for  Collimating 

Spherical  optics:  Point  sources  ("dipoles"  or  wave-guide  apertures)  in  conjunction  with  spherical 

reflectors  or  lenses. 
Cylindrical  optics:   Line  sources   (arrays,  reflectors,  lenses)   in  conjunction  with  cylindrical 

reflectors  or  lenses. 

Arrays  or  Combinations  of  Above  Devices 
Long-wire  antennas. 

Wave  antenna. 

V-antenna. 

Rhombic  antenna,  etc. 
Linear  arrays. 

Broadside  array. 

End-fire  array. 

Fishbone. 

Musa. 

Turnstile. 

Clover  leaf,  etc. 
Curtain  arrays. 

Franklin. 

Pine-tree. 

Chireix-Mesny. 

Sterba. 
Arrays  of:  Horns;  quasi-optical  devices;  wave-guide  apertures. 

RADIATION,  ABSORPTION,  AND  RECIPROCITY.  In  general,  the  radiation  from 
an  antenna  can  be  calculated  if  the  tangential  magnetic  and  electric  fields  subsisting  over 
any  closed  surface  containing  the  antenna  are  known.  The  tangential  magnetic  field 
measures  linear  density  of  an  equivalent  electric  current  in  the  surface  at  right  angles  to 
the  tangential  field;  likewise  the  tangential  electric  field  measures  a  linear  density  of 
"magnetic  current."  General  formulas  (beyond  the  scope  of  this  article)  exist  for  calculat- 
ing from  these  the  radiated  field  (S.  A.  Schelkunoff,  Reference  1,  9.1-7,  9.1-9  and  9.1-10). 
They  are  necessary,  for  example,  in  calculating  the  radiation  from  an  electromagnetic 
horn.  In  many  other  cases,  including  practically  all  antennas  of  older  types  where  the 
energy  prior  to  radiation  was  guided  near  the  outside  of  conductors  rather  than  through 
hollow  guides,  this  procedure  simplifies:  the  closed  surface  now  may  be  taken  as  the  surface 
of  the  conductor  of  the  antenna  itself  and  the  electric  currents  of  the  equivalent  sheet 
become  the  usual  antenna  currents,  while  the  "magnetic  currents"  become  zero  owing  to 
the  disappearance  of  a  tangential  electric  field  at  the  surface  of  the  conductor.  The  fact 
that  in  the  lower-frequency  ranges  the  electric  current  in  conductors  can  be  conveniently 
measured  by  ammeters  gives  a  special  importance  in  those  ranges  to  formulas  in  which 
the  antenna  current  is  assumed  to  be  known. 

An  analogous  process  occurs  in  reception,  in  which  equations  might  be  set  up  for  inte- 
grating the  total  effect  on  the  receiver  due  to  the  electric  and  magnetic  distributions  over 
a  closed  surface  containing  the  antenna,  but,  since  these  fields  are  in  part  reradiation 
associated  with  the  response  of  the  receiving  antenna,  the  problem  is  more  involved  than 
that  of  the  transmitting  antenna  and  it  is  common  to  invoke  the  law  of  reciprocity  instead.  . 
As  applied  to  radio  wave  propagation  through  a  simple  linear  medium  (excluding  non- 
linear circuit  elements  and  the  ionosphere),  this  law  says  that,  if  there  is  a  single  zero- 
impedance  generator  in  the  transmitter  and  a  zero-impedance  ammeter  in  the  receiver, 
the  generator  and  the  ammeter  may  be  interchanged  without  affecting  the  current 
measured.  An  alternative  expression  of  the  law  is :  if  a  constant-current  generator  in  the 
transmitter  produces  a  reading  in  an  infinite-impedance  voltmeter  in  the  receiver,  the 
generator  and  the  voltmeter  may  be  interchanged  without  affecting  the  reading.  These 


PRINCIPLES  OF  LINEAR  CONDUCTOR  ANTENNAS       6-65 

statements  say  nothing  explicitly  about  the  power  transfer,  but  the  following  one  does:  if 
the  internal  resistance  R  of  a  generator  having  zero  reactance  is  matched  to  the  trans- 
mitting antenna,  and  if  the  receiving  antenna  is  matched  to  a  load  whose  impedance  is  R, 
then  the  power  transfer  will  not  be  affected  by  interchanging  the  generator  and  the  load. 
It  follows  from  any  of  these  statements  that  a  given  antenna  has  the  same  directional  pattern 
for  receiving  as  for  transmitting. 


27.  PRINCIPLES  OF  LINEAR  CONDUCTOR  ANTENNAS 

What  takes  place  in  even  a  simple  transmitting  antenna  is  a  matter  of  such  great  com- 
plexity that  a  rigorous  description  is  beyond  the  scope  of  this  article;  nevertheless  it  will 
be  well  to  recognize  the  nature  of  the  problem.  One  aspect  of  it  is  the  calculation  of 
external  fields,  assuming  a  knowledge  of  the  distribution  of  current  in  the  conductors. 


Feeders  all  excited  In  same  phase 
(a)  Sterba  array 


ParasifTc 
'radiator 


To  transmitter 


(5)  Chireix-Mesney  array 


Feeders  all  excited  In  same  phase 


(e)  Pine  Tree  arrays 

th+4   t 

f 

t   t 

t 

t]      1 

* 

fc 
t 

>      {g      {s      js 

IB      is      £      t 

t 

ll  t 

t 

1 

1 

t|      t 

4 

FtankFin  arrays 


To  transmitter 
(/)  RCA  Broadside  arrays 


FIG.  3.    Typical  Antenna  Arrays 

This  problem  is  completely  solved  by  integrating  for  each  point  in  space  the  retarded 
effects  of  currents  at  different  parts  of  the  antenna.  To  the  extent  that  we  are  satisfied 
with  the  assumed  current  distribution,  a  satisfactorily  complete  formal  solution  is  always 
possible.  Many  results  of  great  engineering  usefulness  have  been  so  obtained;  the  usual 
assumption  is  that  the  current  is  distributed  sinusoidally,  that  the  standing-wave  pattern 
is  formed  by  equal  waves  traveling  oppositely  with  the  velocity  of  light  (3  X  10s  meters 
per  second) ,  with  current  nodes  at  open  ends  of  conductors.  This  means  that  the  standing 
wave  will  have  minima  at  intervals  of  one-half  wave,  with  maxima  at  intermediate  points, 
the  instantaneous  current  being  oppositely  directed  either  side  of  a  minimum,  as  shown 
in  Fig.  Id,  Ik,  and  \i.  In  certain  problems  it  is  convenient  to  regard  the  radiation  as 
issuing  from  certain  centers,  such  as  the  midpoint  or  maximum  between  adjacent  minima. 
On  this  view  a  small  radiator  may  have  a  center  of  radiation  much  as  an  extended  body 
has  a  center  of  mass.  For  sections  of  conductor  whose  length  is  small  compared  with  a 
wavelength  the  current  distribution  may  be  substantially  uniform,  as  in  the  vertical  lead 
of  a  large  flat-top  antenna,  or  in  a  loop  small  compared  with  a  wavelength.  There  is  a 
considerable  field  where  the  engineer  may  employ  this  concept  without  apology  to  the 
mathematician . 

There  are  other  cases  where  he  needs  to  watch  his  step  carefully,  as  for  example  in  the 
antenna  shown  in  Fig.  Za  where  for  qualitative  purposes  the  standing  wave  is  shown  along 


6-66 


PASSIVE   CIRCUIT  ELEMENTS 


the  folds  of  the  wire.     The  existence  of  current  minima  in  this  antenna  was  checked 
experimentally  by  E.  J.  Sterba,  but  he  found  that,  as  the  antenna  was  extended  by  adding 

more  sections  at  the  top,  the 

(1)     (2)    (3)   (4)    (5)  (6)      (7)  lower    minima    became    less 

definite  and  ceased  to  be  nulls. 
This  is  an  extreme  case  per- 
haps beyond  mathematical 
solution,  but  it  illustrates  the 
matter. 

A  second  aspect  of  this  the- 
oretical problem  is  the  actual 
determination  of  the  currents 
and  voltages  that  exist  on  an 
antenna  having  specified  the 
applied  or  external  forces. 
S.  A.  Schelkunoff  has  given 
a  general  theory  of  certain 
forms  of  antenna  which  is 


JL 

2 

Characteristic 
impedance 


FIG. 

equa: 

cert 


. 

antenna  of  any  shape  is  similarly  represented,  except  that  the  char- 
acteristic  impedance  is  variable. 


ftrfq;T1orv    trfln*mi««irm    linoc 
ordinary    transmission    lines, 

the  distribution  of  a  linear 
charge  density  is  the  same  as 
that  of  voltage,  but  this  is  not  generally  true  in  antennas.  The  explanation  is  that  antennas, 
unlike  ordinary  transmission  lines,  support  more  than  one  mode  of  propagation.  As  far  as 


.10000 
8000 

6000 
4000 


2000 

.1000 

J  80° 
°  600 

8  400 

03 

.S2 
"t/1 

2  200 

D 
Q. 

_C 

100 
80 
60 
40 

20 
10 


y  soo 


600 


0.20 


0,25 


0.30 


0.35 


0.40 


0.45 


0.50 


FIG.  5.    The  Input  Resistance  of  Hollow  Cylindrical  Antennas  in  Free  Space.    For  vertical  antennas 
over  a  perfectly  conducting  ground  divide  the  ordinates  and  Ka  by  2. 

the  current  associated  with  the  principal  mode  is  concerned  (the  mode  that  we  are  think- 
ing of  when  we  draw  the  oversimplified  current-distribution  curves  which  we  have  just 


PRINCIPLES  OF  LINEAR  CONDUCTOR  ANTENNAS      6-67 


been  discussing),  radiation  is  strictly  an  end  effect:  "It  is  permissible  to  think  that  a  wave 
emerging  from  a  generator  in  the  center  of  an  antenna  is  guided  by  the  antenna  until  it 
reaches  its  boundary  sphere  passing  through  the  ends  of  the  antenna  and  separating  the 
antenna  region  from  external  space;  at  the  boundary  sphere  some  energy  passes  into  ex- 
ternal space  and  some  is  reflected  back — a  situation  existing  at  the  juncture  between  two 
transmission  lines  with  different  characteristic  impedances."  This  will  be  clearer  by  ref- 
erence to  Fig.  4,  taken  from  Schelkunoff,  where  the  "antenna  region"1  or  -sphere  is  re- 
placed by  a  transmission  line  having  length  equal  that  of  the  antenna,  the  line  having 
appropriate  characteristic  impedance  K  and  terminating  impedance  Z*.  The  legend  makes 
the  figure  self-explanatory. 


0.45 


0.50 


FIG.  6.    The  Input  Reactance  of  Hollow  Cylindrical  Antennas  in  Free  Space.    For  vertical  antennas 
over  a  perfectly  conducting  ground  divide  the  ordinates  and  Ka  by  2. 

The  input  resistance  and  reactance  of  perfectly  conducting  cylindrical  antennas  in  free 
space  as  given  by  Schelkunoff  are  plotted  in  Figs.  5  and  6,  the  parameter  Ka,  the  char- 
acteristic impedance,  being  given  in  Fig.  7.  These  curves  were  obtained  on  the  assumption 
that  the  current  flowing  over  the  edge  at  the  top  of  the  antenna  is  zero.  For  Ka  <  700 
there  is  a  small  current  over  the  edge  which,  if  included  in  the  calculations,  would  increase 
the  maxima  of  the  input  resistance  and  reactance  by  a  small  percentage.  Note  the  points 
where  input  reactance  is  zero,  and  the  deviation  of  the  lengths  from  0.25X  and  0.5X,  as 
found  experimentally  by  C.  R.  Englund. 

In  some  low-frequency  antennas  where  radiation  resistance  is  low,  the  input  reactance 
has  often  been  calculated  by  regarding  the  antenna  as  a  transmission  line  without  resist- 
ance, using  the  formula 

X  -  -  -yp  cot  (cozvz^Ji )  CD 

where  Li  and  Ci  are  inductance  and  capacitance  per  unit  length,  I  the  total  length,  and  w 
the  angular  frequency.    Such  a  line  is  resonant  when  I  =  X/4,  3X/4,  etc.,  and  antiresonant 


6-68 


PASSIVE   CIRCUIT  ELEMENTS 


when  I  —  X/2,  X,  etc.    If  a  lumped  inductance  is  connected  in  series  with  the  long  antenna 
assumed,  resonance  will  occur  when 

coZ/  —   Y —  cot  (ool v  L/iCi )  —  0 

The  effect  of  capacitive  loading  may  be  found  in  a  similar  manner. 

When  an  antenna  of  this  kind  does  not  exceed  a  quarter-wave  in  length  it  may  be  roughly 
considered  part  of  a  resonant  circuit  made  up  of  lumped  inductance  and  capacitance  as 


1400 


1000 


800 


600 


200 


1000 


I/a 


10,000 


100,000 


FIG.  7.     The  Average  Characteristic  Impedance:  (1)  Cylindrical  Antenna,  (2)  Spheroidal  Antenna, 
(3)  Antenna  of  Rhombic  Cross-section.    I  is  the  length  from  middle  of  antenna  to  the  ends;  a  is  the 

maximum  radius  of  conductor. 

follows:  Le  —  ZLi/3  henrys  and  Ce  =  IC\  farads  together  with  whatever  loading  may  have 
been  added.    The  shorter  the  antenna  the  better  is  the  approximation. 

RESISTANCE  OF  ACTUAL  ANTENNAS.  Antenna  resistance  is  the  quotient  of  the 
mean  power  supplied  to  the  antenna  divided  by  the  mean  square  of  the  current  referred 
to  a  specified  point  of  the  antenna.  It  thus  includes  a  component  associated  with  the 
useful  radiation  of  power  and  others  related  to  undesirable  losses  in  the  conductors, 

ground,  etc.  Radiation  resistance  is  thus 
the  quotient  of  the  mean  radiated  power 
divided  by  the  mean  square  of  the  cur- 
rent referred  to  the  specified  point,  and 
radiation  efficiency  is  the  ratio  of  radia- 
tion resistance  to  total  resistance. 

Figure  8  shows  the  general  way  in 
which  radiation  resistance,  with  the 
losses  in  the  ground  and  in  the  conduc- 
tors, varies  with  frequency.  An  efficient 
antenna  being  "one  in  which  radiation 
resistance  predominates,  the  desirable 
operating  range  is  well  to  the  right  in 
the  figure.  At  the  lowest  of  radio  fre- 
quencies, however,  economic  factors 
makes  high  efficiencies  unrealizable,  and 
efficiencies  of  a  few  per  cent,  or  less, 
are  representative.  In  the  example 
shown  in  Fig.  8,  the  wire  resistance  is 
substantially  independent  of  frequency, 
but  in  general  this  is  not  true. 

The  radiation  resistance  of  a  straight  vertical  wire  of  infinitesimal  diameter  referred  to 
the  point  where  it  connects  through  a  coupling  or  load  impedance  to  a  perfect  ground 
varies  from  zero  to  36.6  ohms  as  its  length  increases  from  zero  to  a  quarter-wave  (half 
the  value  given  in  Fig.  5) .  A  convenient  approximation  for  the  radiation  resistance  pro- 
vided that  the  actual  height  is  well  below  X/4  is  1607r2A2/X2,  where  h  is  an  effective  vertical 
length  of  the  radiator  and  X  is  the  wavelength  in  the  same  units.  If,  for  example,  we  con- 


Resistance  -  Ohms 
i-»  K>  to  4*  u 

o  o  o  o  o  c 

| 

,    / 

1 

0 

De'sirable 
perating  Rang 

k- 

e-j_ 

7~~ 

/ 

i 

/ 

Y 

i 

j& 

/ 

i 

<^ 

^l« 

/ 

\ 
i 

^ 

^ 

? 

$ 

k\ 

^ 

. 

*$> 

A 

£> 

\ 

\ 
\ 

^ 

\ 

X, 

-C— 

^ 

ll* 

sisti 

J«. 

ofjw 

\ros_ 

—  -  — 

^.-~ 

ix. 

—  - 

Groin 

1.^ 

ssji 

,  — 

— 

)              25             50             75            100          125          15 
Frequency  •  Kilocycles 

FIG.  8.    Components  Which  Go  to  Make  TJp  the  Total 
Resistance  of  a  Simple  Antenna 


PRINCIPLES  OF  LINEAR  CONDUCTOR  ANTENNAS      6-69 


sider  a  vertical  wire  A/4  long  (which  we  have  just  stated  is  too  long  for  this  approximation), 
the  assumed  sinusoidal  current  distribution  makes  h  =  2/x  -  X/4  =  A/2?r,  and  radiation 
resistance  of  40  ohms  is  indicated.  Comparison  with  the  correct  value  of  36.6  ohms 
indicates  that  for  antennas  shorter  than  X/S  the  approximation  will  usually  be  acceptable. 

When  the  operating  frequency  is  low  it  is  uneconomic  to  construct  a  vertical  wire  even 
approaching  a  quarter-wave  in  length.  Under  these  circumstances  it  is  customary  to 
combine  a  rather  large  flat-top  with  a  moderate  vertical  lead,  in  order  to  hold  costs  to 
the  minimum.  This  leads,  however,  to  low  radiation  resistance  and  so  requires  that  other 
resistances  be  kept  correspondingly  low.  Thus,  if  the  radiation  resistance  should  be  as 
low  as  0.05  ohm,  an  elaborate  counterpoise  or  other  ground  system  would  be  necessary  to 
keep  the  losses  within  reasonable 
bounds.  Grounded  antennas 

ANTENNA  IMAGES.  A  portion 
of  the  wave  radiated  from  an  an- 
tenna is  "reflected"  by  the  ear  that 
points  some  distance  away.  To  an 
observer  located  at  a  considerable 
distance  the  total  radiation  appears 
to  be  made  up  of  two  components, 
one  which  arrives  directly  from  the 
antenna  itself  and  the  other  which 
appears  to  be  coming  from  a  virtual 
antenna  located  below  the  surface 
of  the  earth.  The  latter,  sometimes 
known  as  an  image  antenna,  behaves 
in  different  ways  depending  on  the 
soil  adjacent  to  the  antenna.  Figure 
9  pictures  the  relation  between  the 
effective  currents  in  real  and  image 
antenna,  assuming  the  earth  to  be 
perfectly  reflecting. 

FIELDS  ASSOCIATED  WITH  AN  ANTENNA.  The  wave  radiated  from  an  antenna 
appears  as  two  fields,  (1)  an  electric  field  8  which  may  be  measured  in  volts  per  meter, 
and  (2)  a  magnetic  field  which  may  be  specified  in  amperes  per  meter.  These  components 
are  so  inseparable  as  to  be  regarded  as  two  aspects  of  the  same  thing.  They  are  perpen- 
dicular to  each  other  and  to  the  direction  of  wave  propagation  and  at  any  point  of  observa- 
tion are  in  the  same  phase,  reaching  zeros  and  maxima  at  the  same  time.  For  a  funda- 
mental discussion  on  this  subject  the  reader  is  referred  to  Section  5,  articles  26-28. 

The  electric  field  produced  at  a  distance  of  a  few  wavelengths  from  an  antenna  may  be 
calculated  by  means  of  the  general  formula  applying  to  an  elemental  doublet 


FIG.  9.    Images  of  Representative  Forms  of  Antennas 


^~/&I  cos  oj  (t  -  -c 


cos  8 


(2) 


where  8  =  the  field  intensity  of  the  wave  measured  in  volts  per  meter;  /  cos  (cot  +  90°)  = 
current  flowing  in  the  wire  in  amperes;  /  =  frequency  of  the  current  in  cycles  per  second; 
X  —  wavelength  corresponding  to  frequency/;  c«j  =  2-jrf;  t  =  time  in  seconds;  d  =  distance 
to  the  antenna  in  meters;  c  =  velocity  of  light  =  2.998  X  108  meters  per  second;  81  =  the 
elementary  length  of  wire  or  doublet  from  which  radiation  takes  place — it  is  measured  in 
the  same  units  as  X;  and  d  =  angle  of  elevation  of  point  at  which  the  field  is  desired  meas- 
ured relative  to  a  plane  perpendicular  to  the  conductor  5L  In  the  formulas  in  this  section, 
any  other  unit  of  length  can  be  used,  provided  it  is  used  consistently  in  8,  X,  d,  c,  A,  etc. 
The  meter,  however,  is  preferred. 

Equations  for  the  fields  radiated  by  antennas  of  various  shapes,  such  as  given  just 
below,  are  obtained  by  integrating  the  above  doublet  expression  over  the  entire  conductor 
system,  having  due  regard  for  the  distribution  of  the  current.  The  fields  at  great  distances 
from  antennas  are  usually  much  less  than  those  calculated  by  these  formulas.  (See 
Section  10.) 

The  voltage  induced  in  an  incremental  length  of  conducting  wire  by  a  passing  wave  may 
be  found  by  either  of  two  methods.  The  one  starting  with  the  electric  field  of  the  wave  is 
used  the  more.  That  component  of  the  length  lying  parallel  to  the  electric  vector  in  the 
wave  front  is  multiplied  by  8  measured  in  volts  per  unit  length  to  give  the  required  emf. 
This  same  voltage  may  also  be  derived  from  the  simple  dynamo  concept  of  the  number 
of  lines  of  magnetic  force  cut  per  second;  it  will  be  realized,  however,  that  this  is  not  an- 
other component  of  induced  voltage  but  merely  another  approach  to  the  same  one  made 
possible  by  the  identities  expressed  by  Maxwell's  equations. 


6-70  PASSIVE   CIRCUIT  ELEMENTS 

The  instantaneous  radiated  power  flowing  through  each  unit  of  area  perpendicular  to  a 
plane  wave  front  may  be  calculated  by  the  expression 

P  =  0.00265s2    watt  per  square  meter  (3) 

(See  also  Sect.  3  art  28  eq.  [16].) 

If  eq.  (3)  is  used  for  calculating  the  power  picked  up  by  a  single  wire,  it  must  be  assumed 
that  power  is  absorbed  from  a  section  of  the  wave  front  extending  to  about  a  quarter  of  a 
wavelength  either  side  of  the  conductor. 

The  field  intensity  at  a  horizontal  distance  d  from  a  vertical  grounded  wire,  length  h, 
carrying  a  uniform  current  I  is  approximately 


8  =  -  -    volts  per  meter  (4) 

etc 

This  applies  to  the  vertical  lead  to  an  antenna  having  a  relatively  large  flat  top,  the  antenna 
being  located  over  a  perfectly  conducting  earth. 

The  corresponding  value  for  a  coil  or  loop  located  in  free  space  and  having  dimensions 
small  compared  with  the  wavelength  is 

e  =*  —  —z  —  ANI    volts  per  meter  (5) 

dc* 

where  N  is  the  number  of  turns  and  A  is  the  area  in  square  meters. 
The  effective  vertical  length  of  a  loop  in  free  space  is  given  by 

2-rrAN 

h  —  —  -  —    meters  (6) 

A 

The  fie.d  intensity  for  a  simple  vertical  quarter-wave  antenna  grounded  at  its  lower  end  is 
approximately 

60/ 

e  —  —  -     volts  per  meter  (7) 

a 

The  effective  vertical  length  of  a  grounded-quarter-wave  antenna  is 

h  =  —  -     meters  (8) 

2iTT 

In  all  these  cases  the  current  is  to  be  measured  at  the  point  in  the  antenna  where  this 
quantity  is  a  maximum. 

For  formulas  for  field  strength  in  terms  of  radiated  power,  see  Section  10,  Articles  20-24 
of  this  handbook. 

Figure  10  gives  the  directional  diagram  of  a  simple  electric  or  magnetic  radiating  ele- 
ment (a  short  wire  or  a  small  loop).  It  is  worth  noting  that  the  same  diagram  applies 
to  both. 

Axis  of  magnetic  or 
electric  radiating  element 

I 

^y^\ 

/   v/ 

\ 

i  _  Equatorial  plane 
~'       of  elements 


I 

FIG.  10.     Directional  Diagram  of  Electric  and  Magnetic  Dipoles.     The  length  of  the  vector  V  is  pro- 
portional to  signal  intensity. 

The  directional  diagrams  for  both  the  quarter-wave  and  half-wave  antennas  as  well 
as  for  an  antenna  0.62  wavelength  long  are  given  in  Fig.  11.  It  is  to  be  noted  that  for  this 
latter  a  pronounced  spurious  lobe  is  formed.  This  continues  to  grow  as  the  antenna  is 
further  lengthened,  thereby  leading  to  a  considerable  amount  of  high-angle  radiation  and 
possibly  also  to  fading.  The  field  laid  down  in  the  horizontal  direction  is  for  a  given  power 
calculated  to  be  a  maximum  when  the  length  of  the  antenna  is  about  0.62  wave.  (See 


PRINCIPLES  OF  DIRECTIVITY 


6-71 


reference  2.)  The  half-wave  antenna  together  with  its  earth  image  is  roughly  equivalent 
to  two  collinear  equiphased  radiators  and  therefore  is  a  special  case  of  arrays  discussed 
below.  Antennas  a  half-wave  or  so  long  have  become  widely  used  in  broadcasting.  See 
"Antennas  for  Medium-frequency  Broadcasting,"  article  31. 


FIG.  11.    Vertical  Plane  Directional  Diagram  of  a  Single  Vertical  Antenna  of  Various  Lengths.    (Only 
one-half  of  the  total  vertical  section  is  shown.) 


28.  PRINCIPLES  OF  DIRECTIVITY 

An  antenna  which  radiates  or  receives  with  uniform  efficiency  within  a  range  of  direc- 
tion is  said  to  be  non-directional  within  those  limits;  when  it  favors  a  given  direction,  on 
the  other  hand,  it  is  directional.  Thus,  a  vertical  wire  is  non-directional  in  azimuth, 
though  it  is  directional  in  elevation.  Ignoring  non-coherent  radiation,  such  as  light,  it 
may  be  said  that,  for  pure  sinusoidal  currents,  a  completely  non-directional  antenna  does 
not  exist.  It  is,  however,  a  useful  concept  to  define  the  directivity  of  actual  antennas  in 
terms  of  this  imaginary  non-directional  source.  Thus,  the  absoltite  gain  is  commonly 
defined  as 

0-£t  (9) 

where  P  is  the  power  flow  per  unit  area  in  the  plane  linearly  polarized  wave  which  the 
antenna  causes  in  a  distant  region  (usually  in  the  direction  of  maximum  radiation),  and 
PO  is  the  power  flow  per  unit  area  which  would  have  been  produced  if  all  the  power  had 
been  radiated  equally  in  all  directions.  Where  gain  is  referred  to  any  other  standard,  it 
will  be  specifically  mentioned. 

One  special  form  of  radiator,  the  current  element  short  in  comparison  with  a  half-wave, 
has  frequently  been  used  for  reference  in  discussing  directivity.  It  is,  in  fact,  as  non- 
directional  as  a  simple  radiator  can  be.  Actually,  however,  it  has  a  power  gain  of  1.5  over 
the  standard  described  in  the  preceding  paragraph. 


6-72  PASSIVE   CIRCUIT  ELEMENTS 

Though  considerable  directivity  can  at  least  in  theory  be  accomplished  with  an  antenna 
whose  largest  dimension  is  small  compared  with  one  wavelength,  it  is  not  far  from  the 
truth  to  say  that  all  practical  high-gain  antennas  depend  primarily  on  having  current 
distributed  over  dimensions  of  several  wavelengths.  This  statement  applies  both  to 
linear-conductor  antennas  and  to  quasi-optical  devices. 

The  requirements  of  directivity  are  various.  It  may  be  necessary  to  limit  antenna 
effectiveness  to  a  certain  azimuth,  to  a  certain  elevation,  or  to  both.  A  narrow  beam  may 
be  required  in  azimuth  coupled  with  a  wider  one  in  elevation,  as  in  some  transatlantic 
antennas  used  in  the  range  of  high  frequencies;  or  a  relatively  sharp  beam  may  be  wanted 
in  elevation  with  little  or  no  directivity  in  azimuth,  as  in  television  broadcasting;  or 
sharpness  may  be  needed  in  azimuth  with  a  specified  variation  of  intensity  in  elevation, 
as  in  the  "cosecant"  antennas  of  radar,  etc. 

The  special  distribution  of  current  needed  for  directivity  may  be  provided  by  arrays 
of  similar  smaller  radiating  elements  or  by  combinations  of  dissimilar  elements.  These 
building  blocks  may  be  half-wave  wires,  loops,  fiat-top  antennas,  and  so  forth.  The  dis- 
tribution may  be  continuous:  for  example,  an  electromagnetic  horn  may  be  thought  of 
as  an  array  of  an  infinite  number  of  infinitesimal  radiators. 

An  almost  unlimited  number  of  combinations  is  possible,  since  the  directive  effects 
produced  depend  not  only  on  the  relative  positions  and  spacings  of  the  various  units  but 
also  on  the  amplitudes  and  phases  of  their  currents  as  well.  (See  reference  3.) 

Perhaps  the  best-known  arrangement  is  that  of  a  number  of  identical  parallel  antennas 
arrayed  laterally  along  a  straight  line.  Usually  the  element  antennas  carry  equal  in- 
phase  currents.  This  produces  the  strongest  signal  in  a  direction  perpendicular  to  the 
line  of  the  array  and  is  thus  known  as  a  broadside  array.  An  example  is  shown  in  Fig.  11, 
above.  In  some  cases  the  phasing  of  the  currents  is  progressively  delayed  from  antenna 
to  antenna  to  correspond  exactly  with  the  delay  in  that  direction  due  to  finite  wave 
velocity.  Such  an  antenna  is  called  an  end-fire  array  because  the  radiation  is  most  intense 
along  the  line  of  the  array. 

EFFECTIVE  AREA  OF  ANTENNAS.  A  wave  incident  upon  a  receiving  antenna  may 
be  thought  of  as  a  stream  of  energy  possessing  a  certain  power  per  unit  of  cross-sectional 
area.  If  the  receiver  load  is  coupled  to  the  antenna  so  as  to  abstract  the  maximum  power 
available,  then  the  ratio  of  this  maximum  power  to  the  power  incident  on  the  antenna 
per  unit  area  is  defined  as  the  effective  area  of  the  receiving  antenna.  A  somewhat  over- 
simplified view  is  that  the  antenna  presents  this  area  to  the  energy  stream  and  canalizes 
the  corresponding  power  flow  into  the  receiver.  An  excellent  treatment  of  relations  to 
be  discussed  in  this  section  will  be  found  in  an  article  by  H.  T.  Friis  and  W.  D.  Lewis. 
(See  reference  4.) 

The  effective  area  of  a  receiving  antenna  being  by  definition  proportional  to  its  power 
gain,  we  may  make  the  general  statement  that  the  ratio  power-gain  divided  by  effective 
area  has  the  same  constant  value  for  all  antennas.  Numerically  it  turns  out  that  this 
ratio  is  : 


_  _ 
Eff.gain       Aefl.       X' 

an  important  relation  in  antenna  theory.  It  can  be  shown  that  it"  has  a  useful  interpreta- 
tion in  transmitting  as  well  as  in  receiving:  this  same  area  then  measures  that  broadside 
"uniformly  excited"  area  which  would  give  the  same  transmitting  gain  as  does  the  actual 
transmitting  antenna,  the  excitation  being  unidirectional  (e.g.,  as  when  a  reflector  is  used) 
and  the  dimensions  of  the  area  being  large  compared  with  a  wavelength.  Hence,  a  lossless 
transmitting  antenna  in  which  the  radiation  is  associated  with  a  large  uniformly  excited 
area  would,  as  a  receiving  antenna,  make  available  to  the  receiver  all  the  energy  intercepted 
by  its  actual  area. 

A  very  useful  and  simple  free  space  transmission  law  (see  reference  4)  results  by  applica- 
tion of  these  concepts  to  wave  propagation  between  antennas  of  effective  area  AT  and  AR 
(the  subscripts  refer  to  transmitter  and  receiver).  The  power  delivered  to  the  receiver 
is  then  PR,  which  equals 


Total  power  PT 

-  -  -  -  -  -  — 


-  -  -  -  -  -  -  -  —  -  -  •          •  .efT.  (recever)        -A  —  35     —  ^ 

Area  of  sphere  of  radius  d  47rcP        X2 


(receiver) 


giving 


The  fact  that  the  numerical  constant  turns  out  to  be  unity  recommends  this  formula  to 
the  memory. 


DIRECTIVITY   OF  LINEAR  CONDUCTOR  ANTENNAS      6-73 


Directional  Diagrams 


29.  DIRECTIVITY  OF  LINEAR  CONDUCTOR  ANTENNAS 

ARRAYS.  Of  the  many  directive  patterns  that  may  result  from  the  various  spacings 
and  phasings  of  two  antennas,  two  are  of  especial  interest.  In  the  first  the  two  sources 
are  separated  in  space  by  one-fourth  of  a  wave- 
length, and  in  phase  by  one-fourth  of  a  period. 
This  arrangement,  which  is  sometimes  known 
as  a  unidirectional  couplet,  gives  a  cardioid 
pattern  as  shown  in  Fig.  12a,  where  the  unit 
antennas  are  vertical  half-wave  elements.  As 
compared  with  a  single  element  it  effects  a 
power  gain  of  about  2  (3  db).  In  the  other 
arrangement  the  two  elements  are  spaced  one- 
half  wavelength  and  are  driven  in  phase  (see 
Fig.  125).  This  also  gives  a  theoretical  gain  of 
about  2  (actually  somewhat  greater).  The  two 
arrangements  may  be  combined  as  in  Fig.  12c 
to  give  a  total  gain  of  4  (6  db) .  It  is  conven- 
ient to  regard  the  two  antennas  at  the  rear  as 
reflectors  for  those  ahead.  Directional  effects 
such  as  these  are  used  practically  not  only  to 
increase  signal  in  some  desired  direction  but 
also  to  ininimize  its  interfering  effects  in  others. 
Increased  directivity  may  be  obtained  by 
adding  couplets  to  the  arrangement  shown  in 
Fig.  12c.  The  resulting  increase  is  indicated 
by  Fig.  13.  Although  it  is  often  most  conven- 
ient in  practice  to  utilize  spacings  of  one-half 
wave  in  the  array  front,  any  spacing  up  to 
about  3/4  wave  may  be  used.  For  spacings 
less  than  0.6X  it  is  the  total  length  or  aperture 
of  the  broadside  which  is  the  important  crite- 
rion of  gain,  the  gain  variation  due  to  spacing 
being  inappreciable.  Figure  14  facilitates  de- 
termining the  gain  ratio  of  such  arrays.  The 
aperture  there  referred  to  is  about  one-half 
wave  greater  than  the  number  of  wavelengths 
measured  between  extreme  conductors  of  the 
array.  The  reason  for  this  rule  is  that  the  equivalent  area  of  a  thin  wire  properly  coupled 
to  the  terminal  is  finite  (see  paragraph  following  eq.  [3]) . 

When  an  array  is  formed  by  stacking  similar  units  in  tiers  in  the  vertical  direction, 
added  directivity  is  provided  at  some  angle  from  the  vertical,  commonly  at  90°,  that  is, 

in  the  horizontal  plane.  The 
units  may,  for  example,  be  the 
vertical  elements  previously 
described,  and  if  the  elements 
of  a  broadside  array  are  so  ar- 
ranged the  antenna  resembles  a 
curtain.  For  vertical  elements 
the  improvement  obtained  by 
adding  a  small  number  of  tiers 
or  stacks  is  less  than  that 
achieved  by  the  lateral  arrange- 
ment. (See  reference  5.) 

Simple  but  approximate 
rules  for  unidirectional  broad- 
side arrays  are  as  follows: 

1.  The  gain  ratio  of  a  large 
array  of  vertical  couplets  ex- 
tending both  laterally  and  ver- 
tically (and  including  a  reflect- 
ing curtain)  follows  the  general  rule  for  gain  G  =  4^/X2  or,  in  terms  of  a  short  current 
element  as  standard,  G*  =  8/3-  x^L/X2  =  8.4A/X2,  where  the  effective  length  is  taken  one- 
half  wave  greater  than  the  length  between  extreme  conductors  (see  paragraph  following 


FIG.  12.      Horizontal  Directional  Diagrams, 
(a)    Unidirectional  couplet.       (&)    Two  equi- 
phased  antennas  spaced  one-half  wave-length. 
(c)  Two  equiphased  couplets. 


120 
100 

o80 

0 

560 

c 

— 

Arrangement  of  Array, 
o  o   o  o  o   o  o  o-i- 

I 

c 

S 

„ 

•- 

z& 

s 

*•* 

n 

\ 

s 

** 

I 

J 

*' 

s* 

V 

20 
( 

FIG. 

s* 

T 

^, 

--| 

•-* 

/> 

* 

^. 

„- 

s> 

"" 

.-- 

+**• 

^ 

s 

^«« 

—  - 

--* 

l£ 

^f 

•y 

D          4          8         12        16        20       24        28        32        36       4< 
Number  of  Couplets 

13.     Variation  of  Directivity  with  Number  of  Couplets 
Placed  in  Horizontal  Array 

6-74 


PASSIVE   CIRCUIT  ELEMENTS 


GQ-  [3]),  and  the  height  as  A/2  times  the  number  of  tiers  or  stacks.     This  assumes  no  ap- 
preciable gap  between  tiers.    For  a  single-tier  antenna  Gd  =  10A/X2. 

2.  Doubling  the  length  or  the  height,  or  adding  the  "reflector"  curtain  to  the  front 
curtain,  adds  3  db  to  the  gain.  Note  the  exception  already  indicated,  however,  that  in 
going  from  one  to  two  tiers  the  increase  is  only  about  2  db. 

Arrays  of  this  type  have  achieved  considerable  importance  both  in  long-distance  trans- 
mission using  short  waves  (high  frequencies)  and  in  medium-frequency  broadcasting.  In 
the  former  use  very  considerable  power  gains  have  been  employed,  varying  up  to  more 
than  100.  In  the  latter  the  gain  has  been  valuable  in  extending  coverage,  but  the  most 
important  aim  has  been  the  suppression  of  signals  in  certain  directions  at  night  in  order 

to  avoid  interference  with  other 
stations.  Thus  the  directivity  of 
broadcast  transmitting  antennas 
has  two  aims  essentially  corres- 
ponding to  those  mentioned  in  the 
first  section  with  respect  to  direc- 
tivity in  reception,  viz.,  increase  of 
signal  and  reduction  of  interfer- 
ence. 

The  main  problem  in  medium- 
frequency  broadcasting  which  leads 
to  the  use  of  directional  antennas 
is  the  difficulty  of  giving  local  cov- 
erage without  causing  interference 
at  distances  of  a  few  hundred  to  a 
few  thousand  miles  with  other 
transmitters  using  the  same  fre- 
quency.^  This  long-distance  inter- 
PIG.  14.  Graph  for  Predicting  Directivity  of  Arrays  of  Simple  ference  is  propagated  by  reflection 


80 
a 

g60 

03 

cr 
I40 

20 
C 

x 

xl 

^ 

X* 

^ 

^x 

^ 

^* 

^ 

x^1 

X 

^ 

X 

<S 

)             2              4               6              8            10            12            1^ 
Aperture  of  Array,-  Wave-Lengths 

Half-wave  Antennas.     (Aperture  is  expressed  in  wavelengths 
and  is  one-half  wavelength  greater  than  the  horizontal  dis- 
tance between  the  extreme  outside  antennas.) 


from  the  ionosphere,  and  therefore 
not  only  azimuth  but  also  elevation 
must  be  considered.  Regardless  of 
the  height  of  the  reflecting  layer,  the  azimuth  of  waves  between  two  points  lies  along  the 
great-circle  path  with  considerable  consistency,  and  an  antenna  which  directs  a  minimum 
of  signal  at  all  elevations  in  the  vulnerable  azimuth  is  usually  the  most  desirable  solution. 
Sharp  nulls  in  the  elevational  directive  pattern  are  thus  to  be  avoided  if  possible  in  view 
of  the  variability  in  the  heights  of  the  reflecting  layers.  Often,  however,  a  host  of  practical 
considerations  requires  a  compromise  solution.  (See  reference  6.)  Commonly,  also,  the 
problem  is  complicated  by  the  existence  of  more  than  one  vulnerable  direction.  The 
necessity  of  having  the  correct  relative  phases  in  the  unit  antennas  has  required  the 
development  of  techniques  for  controlling,  measuring,  and  maintaining  phases  in  practical 
installations.  (See  reference  7.) 

DIRECTIONAL  CHARACTERISTICS  OF  LONG  WIRES.  When  the  length  of  a 
wire  carrying  a  high-frequency  current  is  progressively  increased,  it  breaks  up  into  oscillat- 
ing sections  as  was  shown  in  Fig.  K.  This  gives  rise  to  radiation  along  certain  preferred 
directions  in  which  in-phase  components  prevail.  Figure  15  shows  the  directional  patterns 
for  certain  representative  cases.  It  will  be  noted  that  the  lobe  designated  as  No.  1  ear 
becomes  progressively  sharper  and  approaches  the  axis  of  the  wire.  At  the  same  time 
smaller  lobes  designated  as  No.  2,  3,  and  4  ears  are  formed. 

Several  long  wires  each  having  characteristics  of  this  kind  may  be  so  combined  as  to 
give  the  arrangement  as  a  whole  very  useful  directional  properties.  In  general,  this  is 
accomplished  by  choosing  arrangements  that  enhance  the  main  lobe  and  at  the  same  time 
discourage  the  spurious  lobes.  The  so-called  tilted  wire,  folded  wire,  and  rhombic  antennas 
are  based  on  this  principle.  (See  reference  8.) 

EFFECTS  OF  SOIL  AND  TERRAIN  ON  DIRECTIVITY.  The  directive  effects  de- 
scribed above  assume  that  the  array  is  divorced  from  any  influence  of  the  earth.  In 
practice,  of  course,  this  is  not  true.  If  the  earth  were  perfectly  conducting  the  array  could 
be  so  elevated  as  to  make  the  image  effect  add  to  that  of  the  array,  thereby  giving  added 
gain  and  a  maximum  intensity  along  the  surface  of  the  earth.  These  ideal  conditions  are 
seldom  attained  in  practice.  At  the  frequencies  at  which  directive  antennas  are  most 
used,  there  is  a  substantial  refractive  effect  in  addition  to  absorption  that  together  tend 
to  distort  the  vertical  directive  characteristic. 

Figure  16  shows  the  calculated  directional  distortion  imposed  by  imperfectness  of  earth 
conductivity  on  both  a  horizontal  and  a  vertical  receiving  doublet  for  a  representative 


DIRECTIVITY  OF  LINEAR  CONDUCTOR  ANTENNAS      6-75 

case.  It  is  to  be  noted  that  the  effect  is  less  marked  with  a  horizontal  half-wave  antenna 
than  with  a  vertical  half-wave,  and  in  both  forms  it  is  such  as  to  cut  down  very  materially 
the  intensities  of  waves  along  the  horizontal.  It  fortunately  happens  that  distant  signals 


Length"*  X 


Length  =l|-X 


No.  2  Ear 


No.  1 
Ear 


No.  2  Ear         No-  I 
No.  3  EarN  \         Ear 
No.  4  Ear, 


No.  2  Ear,   No.  1 
No.  3  Ear>  \    Ear 
No.  4  Ear,  /  j     / 


Length=.2X  Length  =  5X  Length  -B\ 

FIG.  15.    Directional  Diagrams  of  Isolated  Wires  of  Various  Lengths 

arrive  at  an  appreciable  angle  above  the  horizon,  so  that  such  devices  are  still  very  effec- 
tive. While  this  angle  for  transmission  and  reception  of  short  waves  may,  for  short 
distances,  be  nearly  90°,  for  long  distances  it  varies,  say,  from  30°  for  the  lower  frequencies 
to  very  small  angles  for  the  higher  frequencies.  It  would  appear,  therefore,  that  the  dis- 
tortion of  the  vertical  directive  pattern  caused  by  a  soil  of  finite  resistivity  might  constitute 
a  definite  limitation  in  working  with 

50° 


40 


30 


20 


10 


60°       70°     80°    90°     80°     70°      60°         50° 


40° 


30° 


20° 
10° 


1.2     0.8     0.4      0       0.4     0.8     1.2      1.6 
50°  60°70080:)90CS00700  60°  50°     40° 


very  high-frequency  stations.  It  is 
seen  from  Fig.  16  that  horizontal  an- 
tennas are  inherently  high-angle  de- 
vices and  that  vertical  antennas  may 
also  be  high-angle  devices  except  when 
located  over  a  low-loss  earth. 

Advantages  ranging  up  to  10  db 
(see  reference  9)  have  been  obtained 
by  locating  short-wave  antennas  and 
arrays  at  the  tops  of  sharp  declivities 
or  long  slopes.  These  gams  are  com- 
parable with  those  of  the  arrays  them- 
selves and  are  such  as  may  warrant 
considerable  time  in  the  selection  of 
the  site  of  a  short-wave  radio  station. 
They  may  be  explained  either  as  due 
to  the  antenna  being  in  a  position 
where  the  field  distribution  is  more 
favorable  or  by  saying  that  the  de- 
clivity has  effectively  lowered  the  angle 
of  elevation  of  the  antenna  itself. 

As  the  factors  that  effect  vertical  di- 
rectivity vary  markedly  from  point  to 
point  over  the  country  it  is  difficult 
to  present  any  considerable  number 
of  representative  data  in  the  space  here  available.  However,  both  terrain  and  soil  condi- 
tions are  important  insLantenna  design  and  should  be  considered  when  any  large  expen- 
ditures are  to  be  made. 


30° 


2.0 


~0.4      0      0.4    0.8 
Received  Current 


2.0 


FIG.  16.     Vertical  Directional  Diagrams  of  Horizontal 

and  Vertical  Half-wave  Antennas  as  Influenced  by  Finite 

Conductivity  of  Earth 


6-76 


PASSIVE   CIRCUIT  ELEMENTS 


30.  DIRECTIVITY  OF  QUASI-OPTICAL  ANTENNAS  AND  HORNS 

(See  reference  4) 


THE 

which  a 


HUYGENS  SOURCE.     The  optical  concept  of  wave  propagation  according  to 
wave  front  is  considered  an  array  of  secondary  sources  is  of  great  importance  in 

analyzing  the  behavior  of  quasi- 
optical  antennas.  A  formulation 
consistent  with  fundamental  elec- 
tromagnetics has  been  given  by 
S.  A.  Schelkunoff  (reference  1,  9.1 
and  9.24).  Commonly  its  usefulness 
arises  in  situations  like  that,  for  ex- 
ample, at  the  mouth  of  an  electro- 
magnetic horn,  where  we  have  rea- 
son to  believe  that  the  currents 
represented  by  the  wave  front  pre- 
dominate in  ultimate  effect  over 
currents  elsewhere,  such  as  those  on 
the  outside  of  the  horn.  If  we  know 
the  distribution  of  intensities  ovei 
this  aperture,  Huygens'  principle 
can  be  applied.  In  general,  over 
this  surface,  polarization  is  distri- 
buted in  both  the  x  and  y  direc- 
tions, but  for  simplicity  we  write 
the  equation  for  only  the  first  of 
these.  Assuming  the  dimensions  of 
the  aperture  great  enough  to  give 
sharp  directivity,  we  are  mainly 
concerned  with  directions  not  far 
removed  from  the  center  of  the 
main  beam,  which  is  near  the  axis 
of  the  reflector.  Under  these  con- 
ditions the  field  at  a  distance  r  in 
front  of  the  mirror-  is  parallel  to  EQ 
in  the  aperture  and  is  given  by  the 
expression: 


EX.  SHOWN 

AT  7  CM.  A  35  db 

ANTENNA    IS  OBTAINED 

USING  A  64"  PARABOLOID. 

BEAM  WIDTH   IS  I.6*TOTAL  w 

WIDTH  AT  I  db  DOWN. 

PIG.  17.    Nomogram — Paraboloid  Antenna  Data 


dS  being  the  element  of  area  of  the 
aperture,  and  X  the  wavelength. 

THE  APERTURE.  The  theo- 
retical performance  of  a  non-dissi- 
pative  antenna,  which  as  a  trans- 
mitter has  a  uniformly  equal  distri- 
bution of  EQ  over  its  aperture,  is 
useful  for  comparison  with  actual 
transmitting  or  receiving  antennas. 
It  can  be  shown  from  the  above 
equation  (see  reference  10)  that  for 
reception  its  effective  area  equals  the  actual  area  of  its  aperture.  In  other  words,  it  can  be 
expected  to  capture  from  the  wave  all  the  energy  "intercepted."  Note,  however,  that  an- 
tennas in  reception  usually  cannot  do  this,  since  as  a  rule  they  do  not,  in  transmitting,  pro- 
duce a  constant  J?o  over  the  whole  aperture,  and  since,  moreover,  they  may  not  be  large 
enough  compared  with  a  wavelength  to  validate  our  assumptions.  As  applied  to  trans- 
mitting, the  term  effective  area  may  be  interpreted  as  the  area  of  uniform  excitation 
which  would  give  the  same  field  at  the  same  distance  and  with  the  same  total  power  as 
would  the  actual  antenna. 

In  practice  the  effective  area  of  large  apertures  is  usually  considerably  less  than  the 
actual  area,  and  the  ratio  is  an  "efficiency"  factor  which  usually  lies  within  the  range  0.4 
to  0.7,  the  deficiency  being  due  primarily  to  the  non-uniformity  of  intensity  across  the 


DIRECTIVITY   OF  QUASI-OPTICAL  ANTENNAS,   HORNS      6-77 

aperture.  It  should  be  observed  that  requirements  other  than  gain  may  make  it  necessary 
to  avoid  uniformity  across  the  aperture,  such  as  the  desirability  of  suppressing  minor 
lobes  of  the  directional  pattern. 

Other  points  in  connection  with  the  aperture  are:  (1)  whether  or  not  the  amplitude  i« 
uniform,  it  is  usually  important  that  both  phase  and  polarization  be  the  same  at  all  points 
in  a  plane  perpendicular  to  the  desired  direction  of  transmission;  (2)  the  width  of  the 


FIG.  18.    Examples  of  Cylindrical  and  Spherical  Optics 

aperture  a  of  a  broadside  antenna  is  inversely  as  the  angular  beam  width  required  in  the 
plane  containing  that  dimension.  At  the  half-power  points  the  beam  width  is  51  A/a  in 
degrees  for  uniform  illumination  through  a  rectangular  aperture  and  for  non-uniform 
illumination  of  circular  or  elliptical  apertures  it  is  typically  65  A/a.  The  gain  and  beam 
width  of  circular  apertures  having  tapers  of  illumination  commonly  used  at  present  are 
given  in  the  nomogram,  Fig.  17.  When  the  beam  width  required  is  different  in  the  two 
planes,  the  aperture  widths  are  affected  inversely,  an  elliptical  shape  being  common. 
(See  reference  11.) 

Although  the  operation  of  most  microwave  directional  antennas  is  best  understood  in 
terms  of  generation  of  a  plane  wave  front,  there  are  notable  exceptions,  such  as  linear 
end-fire  arrays  and  polyrods. 

POINT  SOURCES  AND  LINE  SOURCES.  In  transmitters  and  receivers,  the  power 
is  conveyed  to  and  from  the  fl.Titp.Trrm  in  transmission  lines  which  are  smaller  than  a 


Reflector 


FIG.  19.    Cylindrical  Collimation 


wavelength  in  cross-section,  whereas  in  the  antenna  the  dimensions  may  be  of  many 
wavelengths.  The  antenna  must,  therefore,  include  a  distribution  system.  This  may 
consist  of  a  branching  system,  such  as  that  used  commonly  in  arrays.  In  most  microwave 
antennas  it  consists  of  a  "primary  feed"  or  radiator  which  launches  a  wave,  this  wave  then 
being  allowed  to  spread  in  azimuth  and  elevation  simultaneously  (spherical  optics),  or 
in  azimuth  and  in  elevation  successively  (cylindrical  optics).  The  latter  two  processes 
are  indicated  in  Fig.  18. 

In  spherical  expansion  the  wave  must  first  be  launched  by  a  primary  feed  antenna  which 
is  basically  a  point  source.  Examples  will  be  given  later.  When,  on  the  other  hand, 
successive  cylindrical  expansion  is  used,  the  wave  from  a  point  source  is  usually  first  con- 


6-78 


PASSIVE   CIRCUIT  ELEMENTS 


fined  between,  closely  spaced  plates  and  allowed  to  expand  in  a  plane,  and  then  the  line 
of  the  wave  front  is  converted  from  a  circular  arc  to  a  segment  of  straight  line;  thereafter, 
the  three-dimensional  wave  front  expands  from  this  line  as  a  cylinder  whose  elements  are 
parallel  to  it.  These  operations  may  be  followed  in  the  example  given  in  Fig.  18  or 
196.  The  line  source  can  be  formed  in  other  ways,  such  as  a  linear  array  of  half -wave 
antennas,  thus  avoiding  the  first  step  of  cylindrical  expansion  (Fig.  19o). 

COLLIMATING  DEVICES:  REFLECTORS  AND  LENSES.  Starting  with  the  energy 
diverging  from  a  point  source,  some  device  is  needed  to  convert  the  wave  front  to  a  plane, 
that  is,  to  make  the  emergent  "rays"  parallel.  For  this  purpose  reflectors  or  lenses  are  used. 

(a)  Parabolic  Reflectors.  The  choice  between  a  paraboloid  and  parabolic  cylinders 
(see  Fig.  18),  depends  on  many  mechanical  and  electrical  considerations.  In  the  past, 
the  paraboloid  has  been  the  more  used.  Among  the  advantages  sometimes  claimed  for  it 
are  greater  electrical  simplicity,  lower  weight  and  better  efficiency,  better  directional 
pattern  in  the  desired  polarization,  and  adaptability  to  conical  lobing  or  spiral  scanning. 


Rear  Feed 


Feed 


(a) 


(6) 


FIG.  20.    Spherical  Collimation 


The  cylinders,  on  the  other  hand,  may  be  simpler  mechanically  and  possess  separate  con- 
trol of  directivity  in  azimuth  and  elevation,  a  point  of  controlling  importance  in  some 
applications. 

For  both  types  of  parabolic  reflectors,  the  feed  is  located  at  or  near  the  focus,  and  the 
section  used  may  be  symmetrical  about  the  axis  or  off  to  one  side  (usually  the  former). 

(b)  Lenses.  Dielectric  lenses  can  be  used,  and  they  have  found  considerable  applica- 
tion in  the  first  step  of  cylindrical  expansion  described  above,  i.e.,  in  the  formation  of 
line  sources,  low-loss  polymers  being  commonly  used  as  dielectric.  For  spherical  optics, 
however,  such  a  lens  becomes  too  massive.  W.  E.  Kock  (see  reference  12)  has  developed 
other  types  of  lenses  particularly  suitable  for  microwave  use,  for  example  that  which 
•employs  metal  plates  instead  of  dielectric  material.  All  lenses  depend  on  having  a  mate- 
rial in  which  the  phase  velocity  is  different  from  that  of  surrounding  space.  If  a  plane 
•wave  were  incident  on  a  bottomless  metallic  honeycomb  in  a  direction  parallel  to  the  cells, 
it  would  pass  through  more  or  less  unimpeded  if  the  frequency  were  above  the  critical 
frequency  of  the  individual  cells  considered  as  wave  guides.  However,  the  phase  velocity 
in  these  wave-guide  cells  would  be  greater  than  that  of  light,  and  the  material  as  a  whole 
would  therefore  possess  a  refractive  index  less  than  unity.  Lenses  can,  therefore,  be  con- 
structed by  grading  the  depth  of  the  cells  in  a  manner  analogous  to  that  of  optical  lens 
practice,  with  this  difference,  that  a  form  which  causes  divergence  in  a  lens  of  glass  will 
produce  convergence  in  a  lens  of  this  cellular  material;  for  example,  a  planoconcave 


DIRECTIVITY  OF  QUASI-OPTICAL  ANTENNAS,   HORNS       6-79 


cellular  lens  can  be  used  as  a  collimator.  If  the  wave  is  linearly  polarized  the  cell  spaces 
may  be  made  indefinitely  wide  in  the  direction  of  the  electric  vector,  the  lens  then  becom- 
ing an  assembly  of  spaced  metallic  plates;  if  in  this  case,  however,  the  electric  vector  is 
made  perpendicular  to  the  plates,  the  wave  passes  through  but  with  substantially  the 
velocity  of  light,  so  that  the  device  does  not  act  as  a  lens.  The  lens  can  be  given  a  stepwise 
reduction  in  thickness  by  means  of  "zoning,"  the  "riser"  of  the  step  being  that  length  of 
wave  guide  necessary  to  include  one  cycle  less  than  a  corresponding  distance  in  free  space. 
(See  Fig.  20.) 

HORNS.  Just  as  the  hollow  wave  guide  is  the  analog  of  the  speaking  tube,  so  the 
electromagnetic  horn  is  the  analog  of  the  acoustical  horn  in  function  as  well  as  in  appear- 
ance. It  is  usually  the  tapered  extension  of  a  metallic  wave  guide,  as  shown  in  Fig.  2a 
and  b  though  it  can  be  excited  in  other  ways. 

As  in  parabolic  reflectors,  the  directional  properties  and  gain  of  horns  are  determined 
by  the  excitation  across  the  aperture,  and  the  considerations  given  at  the  beginning  of 


14 


16 


18 


20  22  24 

Absolute  gain  in  decibels 

FIG.  21.    Optimum  Horn  Data.    The  electric  vector  is  parallel  to  the  dimension 


this  article  on  "The  Aperture"  apply  approximately.  The  distribution  of  intensity  in 
the  aperture  of  horns  tends  to  be  uniform  in  the  E  plane  and  sinusoidal  in  the  H,  a  condi- 
tion resulting  from  the  maintenance  of  the  distribution  natural  to  a  small  wave  guide  in 
passing  from  the  throat  through  the  taper  to  the  aperture.  Roughly  then  the  half-power 
beam  width  in  the  E  plane  of  large  horns  approximates  51  X/aj?,  and  in  the  H  plane 
65  \/an,  provided  that  a,  the  aperture  dimension,  is  larger  than  one  wavelength. 

The  phase  at  the  center  of  the  aperture  of  a  pyramidal  horn  tends  to  lead  that  at  the 
edge  owing  to  the  difference  in  distance  to  the  throat.  For  a  given  size  of  aperture  this 
phase  difference  approaches  zero  as  the  length  is  increased,  and  the  best  length  is  infinity 
since  any  phase  difference  tends  to  reduce  the  gain.  In  a  simple  horn,  therefore,  the  length 
is  more  likely  to  set  a  practical  limit  than  the  aperture,  and  for  a  given  practical  length  it  is 
often  important  to  know  the  flare  or  aperture  that  will  give  greatest  gain.  This  "optimum 
horn"  can  also  be  defined  as  the  minimum  length  of  horn  that  will  give  a  required  power 
gain.  It  does  not  provide  greatest  efficiency  of  aperture  area  but  deliberately  tolerates 
an  increase  in  area  for  a  decrease  in  length. 

Figure  21  gives  the  essential  dimensions  of  optimum  horns,  conical  and  pyramidal, 
which  are  taken  from  the  article  by  A.  P.  King.  (See  reference  13.)  The  area  efficiency 
of  a  large  conical  optimum  horn  is  about  55  per  cent;  that  of  a  very  long  horn  of  the  same 
aperture  is  near  80  per  cent.  (See  reference  13.) 


6-80 


PASSIVE   CIRCUIT  ELEMENTS 


Among  the  forms  of  horn  are  the  conical,  biconical,  sectoral,  and  pyramidal.  Horns 
can  be  used  with  lenses  to  avoid  the  undesirable  phase  difference  across  the  aperture. 
Solid  or  metal  plate  lenses  may  be  used.  The  advantage  of  a  lens  is  that  it  very  greatly 
reduces  the  length  of  the  horn  for  a  given  aperture.  (See  Fig.  20.) 


31.  PRACTICAL  ANTENNA  SYSTEMS 

In  modern  radio  practice  the  highest  frequencies  are  more  than  one  million  times  as 
great  as  the  lowest,  extending  as  they  do  beyond  the  range  from  30,000  cycles  to  30,000 
megacycles.  We  have  emphasized  the  unity  of  principle  within  this  gamut  of  frequency. 
On  the  other  hand,  in  covering  the  field  of  practical  antennas  it  is  necessary  to  examine 
types  in  great  diversity.  Space  prevents  any  attempt  to  be  comprehensive.  In  this  article 
we  can  consider  only  representative  antennas  which  illustrate  different  engineering  prob- 
lems to  be  met  in  practice. 

ANTENNAS  FOR  LOW  FREQUENCIES.  In  Fig,  le  and/  are  shown  prototypes  of 
antennas  important  during  the  first  two  decades  of  "wireless,"  antennas  which  still  are  of 
practical  importance  below  1000  kc.  Their  common  characteristic  is  the  use  of  a  flat-top 


Total  Resistance,  0.4  OJuns    Capacity,  0.053  Mf. 
Radiation  Resistance,  CK05  Ohms  at  18^2  KG. 


FIG.  22.    Typical  Multiple-tuned  Antenna 

and  a  down-lead,  but  these  have  taken  on  many  special  forms — L,  T,  umbrella — with 
single  or  multiple  down-leads.  Except  in  favored  situations  (e.g.,  on  board  a  ship),  it  has 
been  necessary  to  build  more  or  less  elaborate  ground  systems  comprising  buried  wires  or 
an  overhead  "counterpoise"  in  order  to  raise  the  radiation  efficiency  to  an  acceptable  value, 
and  even  then  this  efficiency  might  at  the  lowest  frequencies  be  only  a  few  per  cent.  Two 
general  methods  of  improving  efficiency  are,  first,  to  increase  the  height  of  the  towers 
used  (increase  radiation  resistance),  and  second,  to  use  ground  systems  as  already  stated, 
frequently  accompanied  by  multiple  tuning  (decrease  the  ground  losses) .  Figure  22  shows 
an  Alexanderson  multiple-tuned  antenna  having  six  multipled  down-leads  and  six  tuning 
coils;  its  effectiveness  depends  on  the  fact  that,  although  the  ground  resistances  associated 
with  the  various  down-leads  act  as  though  in  parallel  to  give  a  low  resultant  ground  resist- 
ance, radiation  resistance  is  not  correspondingly  reduced.  Obviously  structures  built  on 
so  large  a  scale  are  very  expensive  and  have  to  be  designed  with  great  attention  to  economic 
factors.  (See  reference  14.) 

Figure  23  shows  an  antenna  installation  for  a  1,50-kc  land  station  such  as  has  been  used 
in  communication  with  ships  at  sea. 


Wire  Rope 


Operati'ng  Frequency  si  50  Kc. 
Radiation  Resistance  el. 6  CO 


FIG.  23.    Flat-top  Antenna  for  Operation  on  150  Kilocycles 


A  wave  antenna  usually  consists  of  a  long  transmission  line  made  up  of  two  wires  spaced 
about  30  in.  and  supported  on  poles  about  25  ft  high  in  accordance  with  standard  pole 
line  construction.  The  length  is  often  about  equal  to  that  of  one  wave.  Although  such 
an  antenna  is  a  relatively  poor  radiator,  its  directional  properties,  together  with  the 
rather  wide  band  of  frequencies  which  it  can  accommodate,  make  it  exceedingly  useful 
in  long-wave  work.  In  particular,  when  used  as  a  receiver,  it  is  able  to  discriminate 
markedly  against  static  arriving  from  other  than  the  preferred  direction  of  reception. 


PRACTICAL  ANTENNA  SYSTEMS 


6-81 


This  provides  a  favorable  ratio  of  signal  to  static.  Because  of  its  broad  frequency  char- 
acteristic it  is  possible  to  attach  two  or  more  receivers  and  simultaneously  receive  several 
frequencies.  These  signals  must,  of  course,  be  arriving  from  the  same  general  direction. 
The  directional  characteristic  depends  among  other  things  on  soil  resistivity.  Such 
antennas  have  not  proved  particularly  effective  in  regions  of  high  rainfall  and  high  con- 
ductivity. Several  wave  antennas  may  be  placed  in  broadside  array  as  described  above 
or  they  may  be  placed  one  back  of  another  and  sidestepped  to  form  a  staggered  array. 

Figure  24  shows  in  schematic  form  a  wave  antenna  and  its  associated  terminating  net- 
work.   The  impedance  Z  is  equal  approximately  to  the  characteristic  impedance  of  the 


Direction  of 


FIG.  24.    Schematic  of  "Wave  Antenna.    (The  length  may  be  as  much  as  a  mile,  the  height  about  25 

feet.) 

antenna.  This  prevents  reflection  and  renders  the  device  essentially  unidirectional.  The 
reflection  transformer  shown  is  an  ingenious  means  whereby  the  accumulated  signal  re- 
ceived between  the  two  wires  and  ground  may  be  transmitted  back  to  a  receiver  located 
at  the  incident  end,  over  the  metallic  circuit  consisting  of  the  two  wires  themselves.  (See 
reference  15.) 

ANTENNAS  FOR  MEDIUM-FREQUENCY  BROADCASTING.  Resonant  antennas 
of  the  general  type  discussed  in  the  previous  sections  (e.g.,  Fig.  23)  have  been  used  hi 
broadcasting.  A  more  common  form,  however,  uses  a  tower  or  mast  itself  as  the  current- 
carrying  conductor  and  radiator.  Some  of  the  forms  which  it  takes,  shown  in  Fig.  25, 
illustrate  its  basic  simplicity.  The  self-supporting  towers  may  be  of  constant  cross-section 
or  tapered  to  a  point  at  the  top.  The  masts,  supported  by  guys  sectionalized  by  insulators 
to  prevent  them  from  taking  part  as  radiators,  are  commonly  tapered  over  the  lower 
fraction  of  their  height  to  a  single  compression  insulator  and  ball-and-socket  joint  at  the 
base.  The  upper  portion  of  the  mast  may  be  tapered  toward  the  top,  but  a  top  without 
this  taper  is  probably  more  common.  (See  reference  16.) 


(«) 


(c) 


FIG.  25.    Typical  Forms  of  Broadcast  Antenna 


All  the  antennas  shown  in  Fig.  25  are  insulated  at  the  base.  There  is  a  form  called  the 
"shunt-fed  antenna"  (see  reference  16),  however,  in  which  the  tower  is  connected  directly 
to  the  ground  network.  The  feed  wire  is  connected,  not  at  the  base,  but  sufficiently  far 
above  it  to  include  an  appreciable  tower  inductance. 

For  daytime  coverage  in  this  frequency  range  the  desideratum  is  a  strong  field  in  the 
horizontal  direction,  since  waves  leaving  in  an  upward  direction  are  ineffective  either  for 
good  or  for  ill  because  they  are  absorbed  by  he  ionosphere.  Assuming  antenna  resistance 
to  be  confined  to  radiation  and  current  to  be  distributed  sinusoidally,  the  maximum  hori- 
zontal field  theoretically  is  obtainable  when  the  height  of  the  radiator  is  5/s  wavelengths 


6-82 


PASSIVE   CIRCUIT  ELEMENTS 


(225  electrical  degrees,  assuming  that  length  of  wave  on  the  tower  is  the  same  as  in  free 
space).  As  shown  in  Fig.  11,  however,  such  an  antenna  will  have  a  strong  minor  lobe  at 
30°  from  the  vertical,  and  at  night  when  ionospheric  absorption  tends  to  disappear  waves 
may  thus  be  received  strong  enough  to  be  comparable  with  the  ground  wave.  Undesirable 
fading  may  then  be  as  serious  as  insufficient  field  strength  would  be,  particularly  in  the 
upper  half  of  the  daytime  frequency  range.  It  has,  therefore,  become  the  practice  to  use 
radiators  of  that  height  and  current  distribution  which  gives  the  best  compromise  between 
field  strength  and  freedom  from  fading,  and  tower  heights  from  0.53  to  0.55  X  (190°  to  200°) 
are  widely  used.  (See  reference  6.) 

The  considerations  of  directional  pattern  just  described  aim  to  protect  the  station's 
listeners  from  its  own  sky  wave.  At  points  beyond  the  normal  ground-wave  range  are 
receiving  sets  tuned  to  other  broadcasters  operating  on  the  same  frequency,  and  they 
must  be  protected  from  interference.  New  minima  of  signal  strength  in  the  elevation 
plane  must  therefore  be  provided  in  that  azimuth,  and  in  view  of  the  great  distances  of 
the  receivers  the  minima  must  be  aimed  far  from  the  vertical,  typically  in  the  order  of  75°. 
For  this  purpose  arrays  of  antennas  are  common ;  such  arrays  for  broadcasting  have  been 
discussed  above  under  "Directivity  of  Linear  Conductor  Antennas:  Arrays."  The  proper 
design  of  such  an  array  is  a  matter  of  some  complication;  not  only  is  the  accurate  locating 
of  towers  and  the  specifying  of  their  relative  currents  an  exacting  matter  for  calculation 
but  also  the  experimental  realization  of  the  desired  amplitudes  and  phases  in  the  presence 
of  the  large  mutual  impedances  which  exist  between  towers  makes  it  necessary  to  provide 
means  for  accurately  adjusting  measuring  and  maintaining  the  currents  both  in  amplitude 
and  in  phase.  (See  reference  7.) 

SPECIAL  ANTENNAS  FOR  BROADCAST  RECEPTION.  Many  broadcast  receivers 
are  supplied  with  built-in  antennas  which,  although  inefficient  as  compared  with  the 
"custom-built"  antennas  that  are  feasible  in  point-to-point  work,  give  adequate  perform- 
ance, thanks  to  the  high  gains  of  receivers  and  the  surplus  signal  laid  down  most  of  the 
time  by  powerful  transmitters.  In  the  few  cases  where  both  received  field  and  atmos- 
pherics are  below  set  noise  advantage  may  be  taken  of  large  antennas.  If  the  antenna  is 
located  where  the  ratio  of  ambient  noise  to  ambient  signal  is  too  great,  it  can  frequently 
be  relocated  in  a  quieter  spot  remote  from  the  receiver  and  may  be  connected  to  the  re- 
ceiver by  a  shielded  transmission  line.  This  is  sometimes  done  in  apartment  houses,  a 
broad-band  amplifier  then  being  employed  to  permit  distribution  to  a  large  number  of 
users.  In  the  nature  of  the  case  broadcast  reception  of  medium  frequencies  does  not 
ordinarily  permit  the  use  of  much  directivity.  (See  reference  17.) 


^-Reflector  Curtain 


FIG.  26.    Arrangement  of  Conductors  and  Impedance  Matching  Devices  in  a  Sterba  Array 

ANTENNAS  FOR  HIGH  FREQUENCIES  (2500  TO  25,000  KG).  In  this  band  of 
frequencies,  almost  exclusively  used  for  long  distances,  the  wavelength  is  small  enough  to 
make  directivity  feasible.  Formerly  the  majority  of  directional-antenna  installations 
employed  broadside  arrays  of  half-wave  elements  operative  over  a  relatively  narrow  band 


PRACTICAL  ANTENNA  SYSTEMS 


6-83 


of  frequencies.  Many  of  these  have  been  replaced  by  some  form  of  long  wire,  which  gives 
considerable  directivity  at  moderate  cost  and  is  sensibly  aperiodic  so  that  it  may  be  oper- 
ated at  several  frequencies  simultaneously.  Resonant  antennas  still  find  use,  however, 
for  example,  where  space  considerations  do  not  permit  the  more  extended  aperiodic  types. 

Figure  3  shows  schematics  of  various  forms  of  broadside  array.  Several  types  of  curtain 
arrays  were  described  by  C.  S.  Franklin,  two  of  which  are  shown  in  the  figure.  Figure  26 
shows  in  somewhat  greater  detail  the  Sterba  array,  including  feed  lines,  impedance  match- 
ing devices  and  provisions  for  sleet  melting.  (See  reference  5.)  As  will  be  noted  by 
tracing  the  connections,  it  is  possible  without  interrupting  service  to  apply  a  60-cycle 
power  to  this  antenna  for  purposes  of  melting  sleet,  a  provision  found  in  earlier  forms  of 
linear  conductor  antennas  (e.g.,  Fig.  22). 

It  has  already  been  pointed  out  that  the  directional  characteristic  of  a  long  straight  wire 
may  be  used  to  produce  antennas  of  marked  directivity.  Two  forms  are  represented  in 
Fig,  27.  The  one  at  the  left,  called  a  "rhombic"  antenna  (E.  Bruce,  reference  8),  presents 


Counterweigh.! 

FIG.  27.    Alternative  Forms  of  Folded  Wire  Arrays 

an  impedance  to  the  terminal  equipment  which  has  a  constant  resistive  value,  making  it 
suitable  over  a  wide  frequency  range  (typically  2  to  1)  provided  that  the  accompanying 
shift  of  directivity  is  appropriate  to  the  medium  of  propagation;  in  transatlantic  work  it  is 
a  fortunate  circumstance  that  the  higher  frequencies  used  by  day  call  for  a  more  nearly 
horizontal  ray  than  the  lower  frequencies  used  at  night,  making  such  an  antenna  suitable 
for  a  wide  range  of  conditions.  When  the  antenna  is  used  for  transmitting  a  considerable 
power  has  to  be  dissipated  in  the  terminating  resistance.  For  details  of  the  resonant-V 
antenna  shown  at  the  right  of  Fig.  27,  as  well  as  several  other  interesting  types,  reference 
may  be  made  to  Carter,  Hansell,  and  Lindenblad  (reference  8) .  Long  wires  are  also  used 
in  the  vertical  plane  for  directional  effects,  as  in  the  "imrerted-V  antenna"  shown  in  Fig.  28. 
An  interesting  application  of  array  principles  to  reception  of  transoceanic  telephony  is 
afforded  by  the  Musa  receiving  antenna,  the  arrays  of  which  are  constructed  with  rhombic 
antennas  as  the  units.  (See  reference  18.)  Over  routes  such  as  this,  radio  waves  of  high 
frequency  usually  travel  by  several  paths 
simultaneously  and  arrive  at  different  angles 
above  the  horizontal.  These  components 
arrive  with  unrelated  radio-frequency  phases 
and  even  with  differences  of  time  delay  which 
are  significant  in  the  audio  range  (of  the 
order  of  a  few  tenths  of  a  millisecond). 
The  Musa  antenna  (Multiple-Unit-Steerable- 
Antenna)  takes  advantage  of  the  spread  in 
angle  of  arrival  to  separate  the  component 
waves,  which  may  then  be  used  singly,  or  in 
"diversity  reception,"  or  combined  with  audio  delay  correction,  etc.  Since  the  waves  are 
more  reliably  distinguished  by  differences  in  vertical  rather  than  azimuthal  angle  of 
arrival,  the  unit  antennas  are  arranged  along  the  great-circle  path  rather  than  in  broad- 
side array.  One  antenna  used  commercially  has  16  unit  rhombic  antennas  in  an  array  2 
miles  long.  The  combining  of  units  is  not  done  in  the  radio-frequency  circuits,  but  it  is 
accomplished  at  intermediate  frequency  through  the  medium  of  a  common  beating  oscil- 
lator. A  multiple  system,  of  phase  shifters  permits  the  separation  and  simultaneous  re- 
ception of  the  different  components  provided  that  they  are  sufficiently  different  in  angle 
of  arrival. 


s//y/r/y//f^^ 
FIG.  28.    Inverted-V  Antenna 


6-84 


PASSIVE   CIRCUIT  ELEMENTS 


ANTENNAS  FOR  VERY  HIGH  AND  ULTRA  HIGH  FREQUENCIES.  The  exten- 
sive application  in  point-to-point  and  mobile  services  of  frequencies  above  30  Me  has  led 
to  a  diversity  of  antenna  types.  Figure  29  illustrates  a  few  types  useful  for  vertical  elec- 
trical polarization  and  suitable  for  mounting  on  a  pole.  Except  for  one  insulator  which  is 
specifically  labeled,  all  lines  shown  represent  conductive  material,  insulators  being  omitted 
in  the  interest  of  simple  representation.  The  lower  portion  of  (a)  is  the  coaxial  which 
feeds  the  upper  A/2  section  at  the  middle  in  a  series  connection  (the  whole  feed-line  current 
flows  into  the  antenna) .  The  portion  of  large  diameter  does  not  touch  the  outer  conductor 
of  the  feed  line  except  at  the  top,  an  arrangement  which  tends  to  minimize  waves  standing 
on  the  supporting  pole.  In  Fig.  296  there  is  a  metallic  conection  between  the  top  of  the 
skirt  and  all  adjacent  parts;  the  feed  is  accomplished  by  bringing  the  inner  conductor 
of  the  feed  line  through  a  hole  in  the  outer  one  at  a  point  inside  the  skirt  which  is'  protected 
from  the  weather.  Figure  29c  and  d  indicates  two  J-shaped  antennas  in  which  the  radiating 


(a) 


(6) 


FIG.  29.    Omnidirectional  Antennas  Using  Vertical  Polarization 

section  is  the  upper  half-wave  of  one  of  the  feed  lines  above  the  point  where  the  other 
ends.  (See  reference  19.)  Figure  29e  shows  a  horizontal  cross  of  four  ground  rods  at  the 
top  of  a  large  supporting  cylinder,  in  the  hollow  end  of  which  is  mounted  an  inner  conductor 
extending  above  it  somewhat  less  than  a  quarter  wavelength,  the  point  of  connection  of 
the  coaxial  line  being  such  as  to  match  impedances.  The  section  below  this  connection 
point  provides  a  strong  mechanical  support  for  the  radiating  member  above.  (See  refer- 
ence 20.) 

When  horizontal  polarization  is  used  in  ultra-high-frequency  broadcasting  there  are 
several  antenna  types  which  may  be  considered.  With  this  polarization  it  is  easier 
than  with  vertical  to  increase  the  gain  by  "stacking."  Several  types  are  indicated  in 
Fig.  30. 

The  "turnstile"  antenna  is  shown  in  Fig.  30a.  (See  reference  21.)  Essentially  it  has 
two  half-wave  horizontal  radiating  members  crossed  at  90°  and  phased  in  quadrature. 
It  is  fed  by  a  system  of  transmission  line*.  When  equal  currents  are  used  in  the  two 
radiators,  the  directional  diagram  in  the  horizontal  plane  is  a  circle  deformed  somewhat 
toward  a  square.  The  vertical  separation  between  stacked  elements  is  one-half  wave. 
The  turnstile  antenna  has  been  adapted  for  broad-band  use  by  employment  of  large  con- 
ductors and  careful  attention  to  detail.  A  cross-section  of  such  an  antenna  on  the  Empire 
State  Building  is  shown  in  Fig.  306,  where  the  cigar-shaped  conductors  and  the  adjacent 
central  parts  are  surfaces  of  revolution  about  the  lines  AC  and  BD.  Separate  transmission 
lines  are  provided  at  F  for  each  of  the  four  radiators.  (See  references  21  and  24.) 

Figure  30c  is  an  "Alford  loop,"  which  is  in  the  form  of  a  horizontal  square  the  length  of 
whose  edge  is  a  matter  of  design,  but  which,  for  descriptive  purposes,  may  be  taken  as 
of  the  order  of  one-third  wavelength.  Current  is  supplied  as  shown,  the  currents  in  the 
four  radiating  members  being  equal  in  magnitude  and  alike  in  phase  as  shown  by  the  arrows 
in  the  diagram.  In  stacking  a  vertical  spacing  of  one-half  wave  is  used. 


PRACTICAL  ANTENNA  SYSTEMS 


6-85 


Figure  30d  shows  a  circular  antenna  (see  reference  22)  which  also  is  substantially  a  loop 
antenna.  The  two  circular  radiating  conductors  indicated  are  electrically  broken  at  B  by 
a  parallel-plate  condenser  without  loss  of  mechanical  continuity  and  strength,  the  whole 
assembly  being  capable  of  support  from  point  A.  The  lower  circle  is  broken  at  C,  from 
which  point  the  system  is  fed  in  the  manner  of  the  "folded  dipole,"  the  "electrical  length" 
of  the  circumference  (taking  account  of  the  loading  capacitance  B}  being  one-half  wave. 
Physically  the  circumference  is  less  than  this.  This  loop  is  attached  to  a  vertical  pole  at  A 
and  is  thus  metallically  grounded.  The  pole  is  inside  the  loop.  The  horizontal  directional 


(a) 

lint  of  Support 


(«£) 


Closed  End 


\Tower 


FIG.  30.    Nondirectional  Antennas  Using  Horizontal  Polarization 

pattern  is  elliptical,  the  maximum  difference  in  field  strength  being  somewhat  less  than  2 
db.  When  these  units  are  stacked  the  vertical  spacing  is  one  wavelength. 

The  "cloverleaf"  antenna,  due  to  P.  H.  Smith  (reference  24)  is  shown  in  Fig.  30e.  This 
consists  of  a  slender  tower  (e.g.,  square)  in  the  form  of  a  conventional  structural-steel 
lattice.  Up  the  center  is  a  conductor,  which,  together  with  the  tower  itself,  forms  a  coaxial 
transmission  system.  The  radiating  "leaves"  are  attached  as  shown,  forming  a  composite 
horizontal  loop.  The  length  of  each  of  these  conductors  is  about  0.4  X.  In  stacking,  a 
half-wave  interval  is  used,  and,  because  of  the  resulting  phase  reversal,  a  clockwise  loop 
has  counterclockwise  loops  immediately  above  and  below  it.  Within  a  range  from  88  to 
108  Me  one  antenna  can  be  changed  from  one  frequency  to  another  by  varying  the  vertical 
spacing  between  loops  of  one  standard  size.  The  horizontal  diagram  is  substantially 
circular. 

The  "rocket"  antenna,  described  by  Andrew  Alford  and  shown  in  Fig.  30/,  is  a  vertical 
cylinder,  metallically  closed  at  both  ends  (in  the  form  shown),  but  having  an  open  slot 
along  one  element  of  the  cylinder.  It  is  fed  as  shown  at  the  point  where  the  cylinder  is 
cut  away  by  establishing  a  voltage  across  the  slot.  It  may  be  thought  of  as  a  "lossy" 
wave  guide  supporting  a  transverse-electric  mode,  the  critical  frequency  of  which  is  (1) 


6-86 


PASSIVE  CIRCUIT  ELEMENTS 


considerably  less  than  that  of  the  dominant  mode  of  an  imslotted  cylinder  owing  to  par 
ticipation  of  the  slot  and  of  the  space  outside  the  cylinder  in  the  propagation  of  the  wave 
and  (2)  somewhat  less  than  the  operating  frequency.  The  metallic  ends  produce  a  stand- 
ing-wave pattern  which  gives  an  approximation  to  uniform  vertical  distribution  of  radiating 
current  over  the  outside,  except  near  the  ends.  The  antenna  is  in  external  effect  somewhat 
like  a  vertical  distribution  of  horizontal  loops.  In  stacking  the  units  are  placed  in  close 
proximity.  The  field  varies  some  3  1/2  db  in  different  azimuths.  The  diameter  is  somewhat 
less  than  one-half  wavelength,  and  the  driving-point  impedance  is  tuned  by  folding  the 
edges  of  the  slot  in.  The  "pylon"  is  a  self-supporting  antenna  employing  somewhat  the 
same  electrical  principle. 

A  horizontal  square  loop  employing  an  interesting  coaxial  feed  system  has  been  de 
scribed  (see  reference  23)  and  is  shown  in  Fig.  30g.  The  sides  of  the  square  are  electrically 
180  in  length,  and  adjacent  sides  are  fed  at  each  corner  in  a  pushpull  manner  from  a 
vertical  coaxial  lying  along  the  axis  of  symmetry,  as  shown.  The  "quarter-wave"  sec- 
tions T  between  the  feed  points  and  this  coaxial  also  provide  impedance  transformation 

See  reference  30  concerning  some  other  antennas  of  interest  in  this  frequency  range 
^MICROWAVE  ANTENNAS.     Practical  microwave  antennas  employ  many  component 
devices  in  various  combinations,  but  we  can  here  recognize  only  general  forms  without 
elaboration.    Brevity  may  lead  to  omission  of  types  of  greater  future  importance  than 

those  included,  but  this  is  un- 
avoidable in  a  field  so  young 
and  lusty. 

Devices  used  in  the  method 
of  spherical  optics  are  illus- 
trated in  Fig.  31,  which  de- 
picts two  point  sources,  and 
in  Fig.  20,  which  shows  cor- 
responding methods  of  colli- 
mation  (production  of  parallel 
rays).  Figures  31a  and  206 
show  a  rear  feed  in  which 
the  energy  comes  through  the 
wave  guide  from  the  left  and 
is  emitted  from  the  two  aper- 
tures toward  the  left.  (See 
reference  26.)  Where  appli- 


Wave  guld 


(a.) 

FIG.  31.    Typical  Point  Sources 


leierence  *o.j  wnere  appli- 
cable, rear  feeds  for  reflectors  provide  a  desirable  method  of  mechanical  support  without 
serious  interference  with  the  electrical  functioning  either  of  itself  or  of  the  reflector. 

Conditions  often  dictate  a  "front"  feed.  Wave-guide  apertures  or  horns  are  frequently 
used.  One  of  the  latter  is  illustrated  in  Fig.  316,  a  sectoral  horn  in  which  the  electric  vector 
might  with  appropriate  design  be  either  vertical  or  horizontal.  The  horn  shown  could 
suitably  be  used  with  a  paraboloidal  reflector  having  an  elliptical  or  rectangular  shape  the 
greater  dimension  being  the  left-right  one  (note  that  this  horn  radiates  a  beam  which  is 
considerably  wider  horizontally  than  vertically  if  the  aperture  has  a  vertical  dimension 
greater  than  a  wavelength).  A  wave  guide  whose  open  end  faces  the  reflector  is  often 
used  (with  or  without  a  flare)  particularly  where  the  paraboloidal  reflector  subtends  large 
angles  both  in  elevation  and  azimuth. 

When  the  feed  is  located  in  the  path  of  the  reflected  wave,  as  in  Fig.  206,  part  of  the 
energy  will  re-enter  it  and  travel  back  along  the  transmission  path  into  the  transmitter. 
This  may  be  great  enough  to  cause  trouble.  One  of  the  solutions  for  this  difficulty  is 
indicated  in  Fig.  20a,  where  the  portion  of  the  reflector  which  might  cause  a  wave  to  be 
returned  to  the  feed  is  omitted,  and  the  feed,  which  is  at  the  focus,  is  directed  toward  the 
active  reflecting  surface.  A  similar  solution  in  which  the  path  from  feed  to  reflector  is 
enclosed  in  a  horn  is  indicated  in  Fig.  20c.  Figure  20d  shows  how  the  spherically  expanding 
wave  from  a  point  source  can  be  collimated  by  a  metal  plate  lens,  such  as  has  been  described 
above.  A  lens  in  which  zoning  has  been  introduced  to  reduce  lens  thickness  is  indicated 
in  Fig.  20e,  while  /  suggests  the  advantage  of  a  lens  in  shortening  a  highly  directional  horn. 
•  Figure  19  illustrates  some  principles  of  design  involving  cylindrical  optics.  Figures 
19a  and  196  include  line  sources  employing  a  linear  array  of  dipoles  and  a  sectoral  parabola 
excited  m  the  TEQ1  (rectangular)  mode  respectively.  (Note  that,  according  to  the  accepted 
convention,  the  designation  TE10  of  the  most  common  mode  in  rectangular  wave  guides 
gives  place  to  TE&  when  the  dimension  parallel  to  the  electric  vector  becomes  greater 
than  the  other  dimension  of  the  cross-section.) 

In  some  applications,  such  as  air-borne  radar,  it  may  be  necessary  to  have  in  azimuth 
a  sharp  concentration,  and  in  elevation  a  wide  spread  of  signal  having  intensity  distributed 


DIRECTION  FINDING 


6-87 


according  to  some  definite  function  of  elevation  angle.     When  this  distribution  gives 
uniform  response  along  the  ground  the  antenna  is  termed  a  cosecant  antenna. 


32.  DIRECTION  FINDING 

Almost  every  type  of  directional  antenna  has,  at  some  time  or  other,  been  used  for 
direction  finding,  but  there  are  a  few  types  which  are  of  especial  importance  due  to  their 
widespread  use.  (See  reference  27,)  Among  those  used  in  the  lower  frequencies  are  the 
loop  in  various  forms  and  the  Adcock  antenna.  Microwaves  have  their  own  distinctive 
methods,  such  as  "lobing"  and  "scanning."  The  directivity  may  be  in  the  receiver  or 
in  the  transmitter,  and  in  radar  it  is  commonly  used  in  both  simultaneously. 

The  loop  (see  Fig.  Ifc)  has  great  simplicity  to  recommend  it  but  also  these  objectionable 
features  to  be  avoided:  it  does  not  distinguish  sense  (e.g.,  east  from  west);  it  is  sensitive 
to  so-called  antenna  effect  (response  to  electric  vector  independent  of  direction  of  arrival 
of  the  wave) ;  and  errors  are  found  when  the  arriving  waves  have  a  downward  direction. 

Signals  arriving  in  a  horizontal  direction  induce  emTs  in  a  loop  in  proportion  to  its  area 
and  the  magnetic  field  of  the  wave.  At  long  waves  most  loops  are  small  compared  with  a 
wavelength,  and  therefore  they  tend  to  pick  up  only  small  signals.  This  bona  fide  emf  is 
therefore  subject  to  interference  from  a  spurious  signal  derived  from  the  electric  rather 
than  the  magnetic  field  (a  more  acceptable  statement  would  be  "from  e  rather  than  from 


Break  in  shield - 


(f) 


FIG.  32.    Methods  of  Direction  Finding 


).  Although  loops  ideally  possess  symmetry  which  will  suppress  this  component, 
as  a  practical  matter  it  is  likely  to  be  an  important  source  of  error.  Figure  32a  (see  refer- 
ence 28)  shows  an  application  of  a  method  for  overcoming  this  difficulty  by  the  addition 
of  a  signal  from  an  auxiliary  small  vertical  antenna  to  balance  out  this  antenna  effect,  thus 
obtaining  sharp  nulls.  The  same  circuit  has  provision  by  switching  for  altering  the  phase 
of  the  added  signal  by  90°,  so  that  it  aids  one  of  the  loop  maxima  and  opposes  the  other, 
thus  establishing  the  sense  (east  or  west) .  A  practical  embodiment  of  this  type  of  circuit 
is  shown  in  Fig.  326  (see  reference  28).  It  illustrates  the  important  method  of  using  a 
shielded  loop  to  reduce  the  effect  of  local  induction  fields  and  to  preserve  electric  sym- 
metry. 


6-88  PASSIVE   CIRCUIT  ELEMENTS 

The  third  kind  of  loop  error  occurs  when  the  direction  of  arrival  is  not  horizontal  and 
there  has  been  some  rotation  of  the  plane  of  polarization.  It  is  really  due  to  pick-up  in 
the  horizontal  wires  of  the  loop  and  can  be  combated  by  essentially  getting  rid  of  them. 
The  Adcock  antenna  illustrated  in  Fig.  32c  avoids  this  difficulty  by  eliminating  the  hori- 
zontal members  and  feeding  the  vertical  ones  by  a  shielded  transmission  line.  ' 

Figure  32d  depicts  the  Bellini-Tosi  loop  method  of  direction  finding  in  which  the  loops 
are  fixed.  By  means  of  shielded  transmission  lines  (usually  horizontal  rather  than  as 
shown  in  the  figure)  and  a  goniometer,  the  direction  can  be  determined  in  an  operating 
room  somewhat  removed  from  the  antenna  itself,  which  in  this  case  may  be  a  large  one. 

Figure  32e  illustrates  the  principle  of  "lobing,"  in  which  by  one  of  several  devices  the 
direction  of  beam  can  be  alternated  between  the  positions  of  the  solid  and  the  dotted  lines. 
When  the  antenna  is  oriented  so  that  the  signals  are  equal,  the  intersection  of  the  lobes 
indicates  the  direction.  This  is  the  method  used  in  many  radars. 

Still  another  method  is  scanning,  as  indicated  in  Fig.  32/,  where  the  beam,  preferably 
very  sharp,  is  swept  periodically  through  a  given  range.  The  direction  of  maximum  re- 
sponse gives  the  desired  information. 

33.  MISCELLANEOUS 

Antenna-testing  methods  are  being  studied  by  the  Antenna  Committee  of  the  Institute 
of  Radio  Engineers,  1  East  79  Street,  New  York  21,  N.  Y.  The  report,  when  issued,  will 
represent  a  much  more  complete  document  than  that  issued  by  the  Committee  in  1938. 
See  reference  29.  From  the  same  source  there  is  now  available  "Standards  on  Antennas, 
etc.  Definitions  of  Terms,"  price  75  cents. 

For  information  on  transmission  lines  see  reference  25. 

For  information  on  antennas  for  aircraft  see  reference  31. 

BIBLIOGRAPHY 

GENERAL 

Federal  Communications  Commission,  "Standards  of  Good  Engineering  Practice  Concerning  Standard 
Broadcast  Stations."  Effective  Aug.  1,  1939,  revised  to  June  1,  1944.  For  sale  by  the  Superintendent 
of  Documents,  Washington  25,  D.  C. 

Ladner,  A.  W.,  and  C.  R.  Stoner,  Short  Wave  Wireless  Communications.    John  Wiley,  New  York  (1933). 

Mcllwain,  K.t  and  J.  G.  Brainerd,  High-frequency  Alternating  Currents.    John  Wiley,  New  York  (1939). 

Schelkunoff,  S.  A.,  Electromagnetic  Waves.    Van  Nostrand,  New  York  (1943). 

Terman,  F.  E.,  Radio  Frequency  Engineering,  Chapter  XIV.    McGraw-Hill,  New  York  (1937). 

1.  Schelkunoff,  S.  A.,  Electromagnetic  Waves,  Chapter  XI.    Van  Nostrand,  New  York  (1943). 
Theory  of  Antennas  of  Arbitrary  Size  and  Shape,  Proc.  I.R.E.,  Vol.  29,  No.  9,  pp.  493-521  (Septem- 
ber 1941). 

2.  Ballantine,  Stuart,  On  the  Radiation  Resistance  of  a  Simple  Vertical  Antenna  at  Wave  Lengths 

below  the  Fundamental,  and,   On  the  Optimum  Transmitting  Wave  Length  for  a  Vertical 
Antenna  over  Perfect  Earth,  Proc.  I.R.E.,  Vol.  12,  No.  6,  pp.  823  and  833  (December  1924). 

3.  Foster,  Ronald  M.,  Directive  Diagrams  of  Antenna  Arrays,  B.S.T.J.,  Vol.  V,  No.  2,  p.  292  (April 

1926). 
Southworth,  G."C.,  Certain  Factors  Affecting  the  Gain  of  Directive  Antennas,  Proc.  I.R.E.,  Vol. 

18,  No.  9,  p.  1502  (September  1930). 

Sterba,  E.  J.,  Theoretical  and  Practical  Aspects  of  Directional  Transmitting  Systems,  Proc.  I.R.E., 
Vol.  19,  No.  7,  p.  1184  (July  1931). 

4.  Friis,  H.  T.,  and  W.  D.  Lewis,  Radar  Antennas.    B.  S.  T.  J.,  Vol.  26,  No.  2,  219-317  (April  1947). 

5.  Sterba,  E.  J.,  Theoretical  and  Practical  Aspects  of  Directional  Transmitting  Systems,  Proc.  I.R.E., 

Vol.  19,  No.  7,  pp.  1194-1196  (July  1931). 

On  calculations  of  characteristics  of  arrays,  etc.,  see:  A.  A.  Pistolkors,  The  Radiation  Resistance 
of  Beam  Antennas,  Proc.  I.R.E.,  Vol.  17,  No.  3,  p.  462  (March  1929) ;  K.  Bechmann,  On  the 
Calculation  of  Radiation  Resistance  of  Antenna  and  Antenna  Combinations,  Proc.  I.R.E.,  Vol. 

19,  No.  8,  p.  1471  (August  1931);  P.  S.  Carter,  Circuit  Relations  in  Radiating  Systems  and 
Applications  to  Antenna  Problems,  Proc.  I.R.E.,  Vol.  20,  No.  6,  p.  1004  (June  1932);  and 
Irving  Wolff,  Determination  of  the  Radiating  System  Which  Will  Produce  a  Specified*Direc- 
tional  Characteristic,  Proc.  I.R.E.,  Vol.  25,  No.  5,  p.  630  (May  1937). 

6.  Duttera,  W.  S.,  Some  Factors  in  the  Design  of  Directive  Broadcast  Antenna  Systems,  R.C.A.  Rev., 

VoL  II,  No.  1,  pp.  81-93  (July  1937). 

Brown,  G.  S.,  Directional  Antennas,  Proc.  I.R.E.,  Vol.  25,  No.  1,  pp.  78-145  (January  1937). 
Guy,  Raymond  F.,  The  New  FCC  Regulations,  Electronics,  August  1939,  p.'ll. 

7.  Morrison,  J.  F.,  A  Simple  Method  for  Observing  Current  Amplitude  and  Phase  Relations  in 

Antenna  Arrays,  Proc.  I.R.E.,  Vol.  25,  No.  10,  p.  1310  (October  1937). 

Kear,  F.  G.,  Maintaining  the  Directivity  of  Antenna  Arrays,  Proc.  I.R.E.,  Vol.  22,  No.  7,  p.'  847 
(July  1934). 

8.  Bruce,  E.,  Developments  in  Short-Wave  Directive  Antennas,  Proc.  I.R.E.,  Vol.  19,  No.  8,  p.  1406 

(August  1931). 
Carter,  P.  S.,  C.  W.  Hansell,  and  N.  E.  Lindenblad,  Development  of  Directive  Transmitting 

Antennas  by  R.C.A.  Communications,  Inc.,  Proc.  I.R.E.,  VoL  19,  No.  10,  p.  1773  (October  1931). 
Bruce,  E.,  A.  C.  Beck,  and  L.  R.  Lowry,  Horizontal  Rhombic  Antennas,  Proc.  I.R.E.,  Vol.  23, 

•No.  1,  -p." 24  (January  1935). 
{Harper,  A.  E.,  Rhombic  Antenna  Design.    Van  Nostrand,  New  York  (1941). 


BIBLIOGRAPHY  6-89 

9.  Potter,  R.  K.,  and  H.  T.  Friis,  Some  Effects  of  Topography  and  Ground  on  Short-Wave  Reception, 
Proc.  I.R.E.,  Vol.  20,  No.  4  (April  1932). 

10.  Friis,  H.  T.,  and  W.  D.  Lewis,  Radar  Antennas,  Section  3.3,  of  reference  4,  above. 

11.  Cutler,  C.  C.,  Parabolic  Antenna  Design  for  Microwaves.    Proc.  I.R.E.,  Vol.  35,  No.  11,  1284 

(November  1947). 

12.  Kock,  W.  E.,  Metal  Lens  Antennas,  Proc.  I.R.E.,  Vol.  34,  No.  11,  p.  828  (November  1946). 
•  Kock,  W.  E.,  Metallic  Delay  Lenses,  B.S.T.J.,  Vol.  27,  No.  1,  58  (January  1948). 

13.  Electromagnetic  Horns,  see:  Barrow,  W.  L.,  and  F.  D.  Lewis,  The  Sectoral  Electromagnetic 

Horn,  p.  41,  and  W.  L.  Barrow  and  L.  J.  Chu,  Theory  of  the  Electromagnetic  Horn,  p.  51, 
Proc.  I.R.E.,  Vol.  27T  No.  1  (January  1939) ;  G.  C.  Southworth  and  A.  P.  King,  Metal  Horns  as 
Directive  Receivers  of  Ultra-short  Waves, 'Proc.  I.R.E.,  Vol.  27,  No.  2,  p.  95  (February  1939); 
S.  A.  Schelkunoff,  Electromagnetic  Waves,  Section  9.23  (Electric  Horns),  p.  360,  Van  Nostrand, 
New  York  (1943);  A.  P.  King,  The  Radiation  Characteristics  of  Conical  Horns  (to  be  published). 

14.  Alexanderson,  E.  F.  W.,  Transoceanic  Radio  Communication,  Trans.  AJ.E.E.,  Vol.  38,  1269-1285 

(1919). 
Alexanderson,  E.  F.  W.,  Reoch,  and  Taylor,  The  Electrical  Plant  of  Transocean  Radio  Telegraphy, 

Trans.  AJ.E.E.,  Vol.  42,  707-717  (1923). 
Lindenblad,  N.  and  W.  W.  Brown,  Main  Considerations  in  Antenna  Design,  Proc.  I.R.E.,  Vol. 

14,  291  (June  1926). 

15.  Feldman,  C.  B.,  The  Optical  Behavior  of  the  Ground  for  Short  Radio  Waves,  Proc.  I.R.E.,  Vol. 

21,  No.  6,  p.  764  (June  1933). 

Beverage,  Rice,  and  Kellogg,  The  Wave  Antenna — A  New  Type  of  Highly  Directive  Antenna, 
Trans.  AJ.E.E.,  VoL  42,  215  (1923). 

16.  Gihring,  H.  E.,  and  G.  H.  Brown,  General  Considerations  of  Tower  Antennas  for  Broadcast  Use, 

Proc.  I.R.E.,  Vol.  23,  No.  4,  p.  311  (April  1935). 

Morrison,  J.  F.,  and  P.  H.  Smith,  The  Shunt-Excited  Antenna,  Proc.  I.R.E.,  Vol.  25,  No.  6,  p.  673 
(June  1937). 

17.  Rettenmeyer,  F.  X.,  Radio  Frequency  Distributing  Systems,  Proc.  I.R.E.,  VoL  23,  No.  11,  p. 

1286  (November  1935). 
Wheeler,  H.  A.,  and  V.  E.  Whitman,  The  Design  of  Doublet  Antenna  Systems,  Proc.  I.R.E.,  VoL 

24,  No.  10,  p.  1257  (October  1936). 
Carlson,  W.  L.,  and  V.  D.  Landon,  A  New  Antenna  Kit  Design,  R.C.A.  Rev.,  VoL  II,  No.  1,  p.  60 

(July  1937). 
Landon,  V.  D.,  and  J.  D.  Reid,  A  New  Antenna  System  for  Noise  Reduction,  Proc.  I.R.E.,  VoL 

27,  No.  3,  p.  188  (March  1939). 

18.  Friis,  H.  T.,  C.  B.  Feldman,  and  W.  M.  Sharpless,  The  Determination  of  the  Direction  of  Arrival 

of  Short  Radio  Waves,  Proc.  I.R.E.,  VoL  22,  No.  1,  p.  47  (January  1934). 
Friis,  H.  T.,  and  C.  B.  Feldman,  A  Multiple  Unit  Steerable  Antenna  for  Short-Wave  Reception, 

Proc.  I.R.E.,  Vol.  25,  No.  7,  p.  841  (July  1937). 
Polkinghorn,  F.  A.,  A  Single  Sideband  Musa  Receiving  System  for  Commercial  Operation  on 

Transatlantic  Radio  Telephone  Circuits,  B.S.T.J.,  Vol.  19,  No.  2,  p.  306  (April  1940). 

19.  Tinus,  W.  C.,  Ultra-high  Frequency  Antenna  Terminations  Using  Concentric  Lines,  Electronics, 

VoL  8,  p.  239  (August  1935). 

20.  Brown,  G.  H.,  and  J.  Epstein,  An  Ultra-high-frequency  Antenna  of  Simple  Design,  Communications, 

Vol.  20,  No.  7,  p.  3  (July  1940). 

21.  Brown,  G.  H.,  and  J.  Epstein,  A  Pretuned  Turnstile  Antenna,  Electronics,  VoL  18,  No.  6,  p.  102 
fcj        (June  1945).     (Gives  bibliography.) 

[Lindenblad,  N.  E.,  Television  Transmitting  Antenna  for  Empire  State  Building,  R.C.A.  Rev., 

Vol.  3,  No.  4T  p.  387  (April  1939). 
Carter,  P.  S.,  Simple  Television  Antennas,  R.C.A.  Rev.,  VoL  4,  No.  2,  p.  168  (October  1939). 

22.  Scheldorf,  M.  W.,  FM  Circular  Antenna,  Gen.  Elec.  Rev.,  Vol.  46,  No.  3,  p.  163  (March  1943). 

23.  Kandoian,  A.  G.,  Three  New  Antenna  Tvpes  and  Their  Applications,  I.R.E.,  Waves  and  Electrons, 

Vol.  1,  No.  2,  p.  70 W  (February  1946). 

24.  A  group  of  short  discussions  on  antennas  for  FM  broadcasting,  i.e.,  circular,  broadcast  loops, 

super-turnstile,  and  cloverleaf  antennas,  Communications,  VoL  26,  pp.  40,  58  (April  1946). 

25.  For  information  on  transmission  lines  see  Section  10  in  this  volume  (Wire  Transmission  Lines); 

also  see  W.  S.  Duttera,  New  Coaxial  Conductor  at  WTAM,  Electronics,  March  1939,  p.  30. 
J.  B.  Epperson,  Installation  of  Coaxial  Transmission  Lines,  Part  I,  Electronics,  July  1939,  p.  30, 
August  1939,  p.  31  (Part  II;  E.  J.  Sterba  and  C.  B.  Feldman,  Transmission  Lines  for  Short-wave 
Radio  Systems,  Proc.  I.R.E.,  Vol.  20,  No.  7,  p.  1163  (July  1932) ;  W.  L.  Everitt  and  J.  F.  Byrne, 
Single-wire  Transmission  Lines  for  Short-Wave  Antennas,  Proc.  I.R.E.,  VoL  17,  No.  10,  p.  1840 
(October  1929). 

26.  Cutler,  C.  C.,  Design  of  Parabolic  Antennas.     See  reference  11  above. 

27.  Keen,  R.,  Wireless  Direction  Finding,  3d  Ed.    EifFe  &  Sons,  London  (1938). 

28.  Martin,  H.  B.,  Small  Vessel  Direction  Finding,  R.C.A.  Rev.,  Vol.  II,  No.  lf  p.  69  (July  1937). 

29.  Report  of  the  Antenna  Committee  entitled:  "Standards  on  Antennas,"  to  be  published  by  the 

Institute  of  Radio  Engineers,  1 E  79  Street,  New  York  21,  N.  Y. 

Cutler,  C.  C.,  A.  P.  King,  and  W.  E.  Kock,  Microwave  Antenna  Measurements.    Proc.  I.R.E.t 
VoL  35,  No.  12,  1462  (December  1947). 

30.  Hidetsugu  Yagi,  Beam  Transmission  of  Ultra  Short  Waves,  Proc.  I.R.E.,  VoL  16,  No.  6,  p.  715 

(June  1928). 

Kraus,  John  D.,  The  Corner-Reflector  Antenna,  Proc.  I.R.E.,  VoL  28,  No.  11,  p.  513  (November 
1940). 

31.  Antennas  for  Aircraft:  P.  C.  Sandretto,  Principles  of  Aeronautical  Engineering,  McGraw-HilL. 

New  York  (1942) ;  F,  D.  Bennett,  P.  D.  Coleman,  and  A.  S.  Meier,  The  Design  of  Broad-Band 
Aircraft-Antenna  Systems,  Proc.  I.R.E.,  VoL  33,  No.  10,  p.  671  (October  1945);  Fit.  Lt.  C.  B. 
Bovill,  J.  AJ.E.E.  (London),  VoL  92,  Part  III,  p.  105  (June  1945);  G.  L.  Haller,  Aircraft 
Antennas,  Proc.  I.R.E.,  VoL  30,  No.  8,  p.  357  (August  1942). 


SECTION  7 
VACUUM-TUBE  CIRCUIT  ELEMENTS 


AMPLIFIERS 
^Tf  BY  Lor  E.  BABTON  PAGE 

1.  Class  A  Amplifiers  ...................  03 

2.  Class  B  Amplifiers  ...................  15 

3.  Class  C  Amplifiers  ............  .  ......  24 

4.  Regeneration  and  Its  Prevention  .......  28 

SPECIAL-PURPOSE  AMPLIFIERS 
BY  E.  L.  CLAKK 

5.  Wide-band  Amplifiers  ................  31 

6.  Cathode  Followers  ...................  47 

7.  Grounded-grid  Amplifier  ..............  49 

8.  In-phase  Amplifiers  ..................  50 

9.  Negative-feedback  Amplifiers  ..........  51 

10.  Pulse  Amplifier  ......................  55 


rNTERMEDIATE-FREQUEH'CY 

AMPLIFIES 
BY  CnAjaLES  J.  Erases 

11.  Factors  Affecting  the  Choice  of  Inter- 

mediate Frequency  .................     56 

12.  Narrow-    and     Medium-bandwidth     I-f 

Amplifiers  .........................     58 

13.  Wide-band  I-f  Amplifiers  .............     63 

MODULATORS 
BY  J.  E.  YOUNG 

14.  Types  of  Modulation  .................  71 

15.  Grid  Modulation  .....................  73 

16.  Plate  Modulation  ....................  74 

17.  Comparison  of  Modulation  Systems  ....  75 


DETECTORS 
^xx.  -^T  VEHNON  D.  LAXDON  PAGE 

18.  Square-law  Detection 76 

19.  Linear  Detection 79 

OSCILLATORS 
BY  CASL  C.  CHAMBERS 

20.  Vacuum-tube  Oscillators 83 

21.  Electromechanical  Oscillators 91 

22.  Cavity  Resonators,  by  I.  G.  Wilson  and 

J.  P.  Kinzer 95 

POWER  SUPPLY 
BY  J.  E.  YOUNG 

23.  Receiver  Power  Supply 106 

24.  Transmitter  Power  Supply 108 

25.  Rectifier  Circuits 110 

RADIO  RECEIVERS 
BY  VEBNON  D.  LANDON 

26.  Types  of  Receivers 117 

27.  Fidelity  Characteristics 125 

28.  Random  Noise 127 

RADIO  TRANSMITTERS 

BY  J.  E.  YOUNG 

29.  Intermediate-radio-frequency  Amplifiers  129 

30.  Power  Amplifiers 131 

31.  Audio  Amplifiers 134 

32.  Telegraph  Transmitters 134 

33.  Installation  of  Radio  Transmitters 134 


7-01 


VACUUM-TUBE  CIRCUIT  ELEMENTS 
AMPLIFIERS 

By  Loy  E.  Barton 

An  amplifier  is  a  device  for  increasing  the  energy  associated  with  any  phenomenon 
without  appreciably  altering  its  quality. 

Amplifiers  used  in  communication  circuits  almost  invariably  employ  thermionic  vac- 
uum tubes  as  the  amplifying  elements.  The  vacuum  tube  is  practically  a  pure  resist- 
ance device  at  low  frequencies.  As  the  frequency  increases  to  a  point  where  the  inter- 
electrode  capacity  impedance  becomes  appreciable  with  respect  to  the  internal  resistance 
of  the  tube  a  phase  shift  is  introduced  which  alters  gain  or  amplification. 

At  still  higher  frequencies  the  time  required  for  the  electrons  to  reach  the  plate  is  ap- 
preciable with  respect  to  a  quarter  cycle.  When  this  transit  time  is  about  20  to  30 
electrical  degrees  the  amplifier  characteristics  are  modified  considerably.  These  ampli- 
fiers still  fall  in  one  of  the  three  basic  classes  of  amplifiers. 

A  class  A  amplifier  is  an  amplifier  in  which  the  grid  bias  voltage  permits  a  steady  plate 
current  flow  of  such  a  value  that  the  plate  current  varies  directly  as  the  grid  voltage  for 
the  complete  cycle  of  360  electrical  degrees.  The  resulting  output  voltage  for  an  ideal 
class  A  amplifier  is  an  exact  reproduction  of  the  grid  voltage. 

The  characteristics  of  the  class  A  amplifier  are  low  power  output  with  a  theoretical 
maximum  plate  power  efficiency  of  50  per  cent  and  an  operating  efficiency  of  about  30 
per  cent  or  less  at  full  power  outputs.  The  plate  dissipation  is  maximum  at  zero  out- 
put, and  the  plate  circuit  output  network  may  be  tuned  or  untuned  for  an  undistorted 
output.  The  average  value  of  plate  current  does  not  change  during  the  cycle  so  that 
the  input  plate  power  is  constant. 

A  class  B  amplifier  is  an  amplifier  in  which  the  grid  voltage  permits  essentially  zero 
plate  current  with  no  signal  applied  to  the  grid  and  the  plate  current  is  proportional  to 
the  grid  swing  when  the  grid  swings  in  a  positive  direction  from  the  bias  point  so  that 
plate  current  flows  for  approximately  180  electrical  degrees. 

The  characteristics  of  the  class  B  amplifier  are  comparatively  high  power  output  with 
a  theoretical  maximum  plate  efficiency  of  78.5  per  cent;  in  practice  the  efficiency  ap- 
proaches 65  per  cent  at  full  output.  The  plate  dissipation  is  a  minimum  and  is  com- 
paratively low  at  zero  signal,  increasing  rapidly  to  approximately  constant  value  at  about 
25  per  cent  full  output.  However,  the  power  input  to  the  plate  increases,  with  signal 
and  power  output,  until  the  peak  output  is  reached.  Therefore,  the  plate  current  is  a 
variable  and  the  plate  supply  voltage  should  have  good  regulation.  The  class  B  am- 
plifier may  be  used  as  a  single-tube  tuned-plate  circuit  amplifier  or  the  plate  circuit  may 
be  untuned  provided  two  tubes  are  used  in  a  pushpull  manner  with  appropriate  input 
and  output  transformers. 

A  class  BI  amplifier  is  an  amplifier  biased  and  operated  as  a  normal  class  B  amplifier 
except  that  the  grid  swing  does  not  go  into  the  positive  region. 

In  general  such  an  amplifier  uses  low-mu  tubes  in  order  that  high  plate  currents  may 
be  reached  without  driving  the  grid  into  the  grid-current  region.  The  efficiency  of  this 
amplifier  is  lower  than  that  of  an  amplifier  in  which  the  grid  is  driven  to  plate-current 
saturation,  and  the  power  output  is  lower.  However,  the  grids  take  no  power  so  that 
input  circuit  losses  and  distortion  may  be  low. 

A  class  ABi  amplifier  is  an  amplifier  so  biased  that  the  pushpull  ^tubes  act  as  a  class  A 
amplifier  for  low  grid  swings  and  go  into  class  BI  operation  at  higher  grid  swings.  The 
grids  are  not  driven  positive. 

The  only  advantage  of  this  type  of  operation  is  that  somewhat  higher  outputs  may  be 
obtained  for  a  given  plate  dissipation  than  can  be  obtained  for  class  A  operation  and  a 
cathode  resistor  can  be  used  for  self-bias.  However,  distortion  is  comparatively  high. 

A  class-  AB2  amplifier  is  similar  to  the  class  ABi  amplifier  except  that  the  grids  are 
driven  into  the  positive  region  for  higher  outputs. 

The  characteristics  of  the  class  AB2  amplifier  are  much  the  same  as  those  of  the  class 
ABi  except  that  the  power  output  is  higher,  and  the  grid  driving  problem  is  about  equal 
to  that  of  the  class  B  amplifier. 

7-02 


CLASS  A  AMPLIFIERS  7-03 

A  class  C  amplifier  is  an  amplifier  in  which  the  grid  bias  voltage  is  appreciably  higher 
than  the  bias  required  for  plate-current  cutoff,  and  the  plate  current  flows  for  a  period 
less  than  180  electrical  degrees  during  the  half  cycle  when  the  grid  swing  is  positive  with 
respect  to  the  bias  voltage.  The  grid  swing  is  usually  to  the  point  of  plate-current  satu- 
ration, in  which  case  the  rms  plate  current  is  proportional  to  plate  voltage  and  is  not 
proportional  to  the  grid  voltage. 

The  characteristics  of  the  class  C  amplifier  are  high  power  output  and  an  average  plate 
efficiency  in  practice  of  70  to  75  per  cent,  which  may  reach  85  to  90  per  cent  under  special* 
conditions.  (See  Fig.  35,  p.  7-27,  and  discussion.)  The  theoretical  maximum  effi- 
ciency for  the  class  C  amplifier  is  100  per  cent. 

1.  CLASS  A  AMPLIFIERS 

GENERAL  USE.  The  class  A  amplifier  is  used  for  audio-frequency  and  radio- 
frequency  voltage  amplification,  principally  because  the  output  voltage  is  a  direct  func- 
tion of  the  input  voltage.  The  amplifying  action  of  the  vacuum  tube  follows  from  the 
equations  developed  for  its  plate  current  in  Section  5.  If  all  the  terms  but  the  first  in 
eq.  (17),  p.  5-43,  are  negligible,  the  voltage  drop  due  to  the  plate  current  of  the  vacuum 
tube  flowing  through  an  external  impedance  is 


or  for  a  particular  component  of  periodicity  &m 

E,  =  -2^=-  (la) 

Tp  +  Zm 

This  voltage  is  a  magnified  replica  of  the  grid  voltage  and  is  said  to  be  amplified. 

The  voltage  amplification  is  the  ratio  of  the  change  in  voltage  across  the  external  load 
to  the  change  in  input  voltage,  or 


V.A.  =     /  *•».      =      =  (2) 

V(r  +  rp)2  +  xm*       *» 

if  there  is  no  drop  in  the  external  grid  circuit.  Two  special  cases  are  of  interest:  First, 
when  the  load  is  a  pure  resistance;  V.A.  =  pr/(rp  -j-  r}  which  will  equal  0.9  jtt  when 
T  —  9rp.  Second,  when  the  load  is  a  pure  reactance,  V.A.  ~  tucm/  V  rp2  -{-  xm*,  which 
will  approximate  0.9  JJL  when  xm  =  2rp. 

The  power  amplification  is  the  ratio  of  the  power  delivered  by  the  output  circuit  to 
the  power  supplied  to  the  input  circuit,  or 


l(rP  +  r)2  -f  xJKn  +  re} 

where  z"  is  the  total  impedance  in  the  grid  circuit. 

For  small  power  outputs,  the  power  amplification  is  very  high  and  is  usually  limited 
only  by  the  input  circuit  losses. 

Cascade  Amplifiers.  The  limitations  of  voltage  and  power  amplification  obtainable 
with  one  tube  make  it  frequently  desirable  to  use  two  or  more  tubes  in  cascade  by  coup- 
ling the  plate  circuit  of  the  first  tube  to  the  grid  circuit  of  the  next  tube,  etc.  Each  tube 
with  its  associated  circuits  is  called  a  stage  of  the  amplifier,  and  the  whole  is  termed  a 
multistage  or  cascade  amplifier.  Cascade  amplifiers  are  usually  classified  by  the  method 
of  coupling  used. 

The  amplification  per  stage  is  defined  as  the  ratio  of  the  grid  voltage  of  one  tube  to  the 
grid  voltage  of  the  preceding  tube.  This  is  frequently  expressed  in  decibels. 

For  the  resistance-coupled  amplifier  shown  in  Fig.  1,  it  is 

V.A.  •.  --  -  -  (4) 

rpTj*       rp  TyXjz 

—  ;  —  {  ---  r  1  —  3  -  r~ 

2t22  T  Zi? 

For  low  frequencies,  and  when  the  grid  biasing  voltage  is  always  negative,  the  amplifi- 
cation limit  per  stage  of  this  type  of  amplifier  is  ft. 

^- 


7-04 


VACUUM-TUBE   CIRCUIT  ELEMENTS 


A  direct  resistance  coupled  amplifier  of  the  type  shown  in  Fig.  1  is  used  when  it  is 
necessary  that  a  d-c  voltage  be  amplified.  It  will  be  noted  that  the  required  B  voltage 
increases  with  each  stage  and  that  independent  filament  supplies  are  needed  for  direct- 
heated  tubes.  In  the  case  of  indirectly  heated  cathodes  a  separate  heater  supply  is  usu- 
ally needed  because  of  the  excessive  heater  cathode  potential. 


FIG.  1.     Direct" Resistance  Coupled  Amplifier,  Common  B  and  C  Batteries,  Separate  A  Batteries 


The  amplification  per  stage  of  the  resistance-capacitance  coupled  amplifier  shown  in 
Fig.  2  is 


V.A. 


zc  +  rp       rpzc       rpZc   ,rP 
I  ---  1  --  ~t  ---  1  ---  r  J- 


(6) 


in  which  zc  is  the  reactance  of  the  coupling  condenser  C.     A  reactance  may  be  substituted 
for  either  r  or  TK,  or  both,  in  which  case  these  constants  are  changed  from  r  to  z  in  eq.  (6)  . 


PIG.  2.    Resistance-capacitance  Coupling 

If  the  input  impedance  to  the  second  tube  is  very  high  and  the  impedance  of  the  coupling 
condenser  negligible  eq.  (6)  reduces  to 

VA.  = £— -  (6a) 


from  which  it  is  seen  that  the  higher  the  resistance  of  the  parallel  combination  of  r  and 
TK.  the  greater  the  realized  amplification  will  be.  The  limiting  value  of  TK.  is  usually 
1  megohm  or  less,  so  that  it  is  of  little  value  to  make  r  greater  than  75,000  to  100,000  ohms. 

The  plate  voltage  supply  must  be  great  enough  to  operate  the  tube  at  a  desirable  point 
and  to  supply  the  drop  in  the  load  resistor  in  order  to  obtain  maximum  voltage  in  the 
succeeding  tube.  A  lower  plate  supply  voltage  may  be  used  with  a  corresponding  re- 
duction of  bias  if  maximum  voltage  output  is  not  required.  This  will  have  little  effect 
on  the  amount  of  amplification  obtained. 

Cascade  operation  of  two  or  more  resistance-capacitance  coupled  audio  amplifier  stages 
is  subject  to  a  low-frequency  oscillation  or  "motor-boating,"  resulting  from  a  feedback 
of  low  frequencies  due  to  the  common  a-c  impedance  of  the  plate  supply  at  frequencies 
so  low  that  the  filter  condensers  are  not  effective.  To  correct  the  motor-boating  diffi- 


CLASS  A  AMPLIFIERS 


7-05 


culty,  it  is  usually  necessary  to  limit  the  low-frequency  response  or  isolate  one  or  more 
of  the  resistance  coupled  stages.  The  low-frequency  response  is  most  easily  reduced  by 
decreasing  the  size  of  the  coupling  condenser  (C  in  Fig.  2). 

A  transformer-coupled  amplifier  is  shown  in  Fig.  3  and  its  equivalent  circuit  in  Fig.  4. 
In  the  equivalent  circuit,  account  is  taken  of  the  distributed  capacitances  (Cj  and  Cg) 


FIG.  3.     Transformer  Coupled  Amplifier 


FIG.  4.     Equivalent  Circuit  of  the  Transformer  Coupled  Amplifier  of  Fig.  3 

of  the  windings  and  of  the  distributed  capacitance  (CM)  between  windings.  The  trans- 
former windings  are  assumed  to  be  poled  so  that  the  effect  of  this  latter  is  a  minimum. 
The  voltage  amplification  of  each  stage  is  then 

V.A.  = 


where 


__ 


,    zs 


—  J- 


Zx  = 


23  H-  24  —  j2t»M  — 


-  3 


and 


Z3  w^ 

Note  that  if  the  incidental  resistances  of  the  transformer  windings  are  neglected  ZM* 
Zx,  and  zy  are  all  pure  imaginaries. 

The  maximum  value  of  V.A.  with  respect  to  0-5  occurs  under  two  conditions:  one  value 
(infinity)  will  give  maximum  value  of  V.A.  for  all  frequencies;  the  other  value,  dependent 
in  a  complicated  manner  on  the  mesh  parameters,  will  give  a  maximum  for  only  one 
particular  frequency.  The  same  result  is  found  when  x^  is  considered  the  variable. 
These  correspond  to  the  cases  of  tuned  and  untuned  transformers.  (See  p.  6-08.) 


7-06 


VACUUM-TUBE   CIRCUIT  ELEMENTS 


When  substantially  equal  amplification  is  required  over  a  broad  frequency  range  the 
distributed  capacitance  should  be  made  as  small  as  possible.  Assuming  xz  and  #5  infi- 
nite and  CM  zero,  eq.  (7)  reduces  to 


24)  +  u*M* 

By  tuning  primary  and  secondary  circuits  as  shown  in  Fig.  5,  the  maximum  amplifi- 
cation (at  one  frequency  only)  is 


VA.  = 


7-4) 


(76) 


A 

D 

^ff 

^!La 

A 

D    ) 

< 

< 

?  s 

C2 

ilililil 

Out 

< 

I|I|I|I| 

i 

^ 

-if 

illhiilll 

FIG.  5.     Tuned  Transformer  Coupled  Amplifier 


Output 


FIG.  6.     Tuned  Coupling.     The  tuned  circuit  is  La-Lb-C.    The  coil  Zk  and  the  capacitor  Ck  are 
inserted  to  get  proper  bias  on  the  grid  tube  2. 


A1 


'  *~d 

5  ^rc»  j?f  > 

0 

Outf 

i 

Tl 

-ill! 

_^ 

-S 

-1 

r~ 

lidhlil— 

FIG.  7.    Tuned  Coupling.     The  tuned  circuit  is  Ca-Ct>-L.    The  coil  Zk  is  inserted  to  get  proper 
potential  on  the  plate  of  the  first  tube. 

Other  forms  of  tuned  amplifiers  are  shown  in  Figs.  6  and  7.     For  the  resonant  fre- 
quency the  amplification  of  the  circuit  of  Fig.  7  is 


VA.  = 


where  z 


fi  -  *r 

\Z          2i2 


(8) 


CLASS  A  AMPLIFIEES 


7-07 


Performance.  Certain  tube  characteristics  are  supplied  by  the  tube  manufacturers 
(see  Section  4)  which  are  used  to  predict  the  performance  of  tubes  as  class  A  amplifiers; 
representative  curves  and  tube  constants  will  be  given  herein  with  sample  calculations  of 
performance.  A  typical  general-purpose  triode  is  the  56-type  tube  whose  plate  char- 
acteristics are  given  in  Fig.  8. 
It  will  be  noted  that  each 
curve  of  plate  current  against 
plate  voltage  is  drawn  for  a 
given  grid  or  bias  voltage 
denoted  as  Ec.  From  these 
curves,  the  various  constants 
of  the  tube  may  be  calculated 
as  explained  in  the  section 
on  vacuum  tubes.  The  con- 
stants for  the  56,  as  calculated 
from  Fig.  8,  at  approximately 
238ftvolts  at  the  plate,  13.5 
volts  bias,  and  5  ma  plate  cur- 
rent, are  plate  resistance  (rp) 
equals  10,000  ohms,  amplifica- 
tion factor  (AI)  equals  15,  and 
the  grid  plate  transconductance 
equals  1500  micromhos. 


200  300 

Plate  Vjolts 


400 


500 


FIG.  8.    Type  56,  Plate  Characteristics 


INPUT  CIRCUIT  CALCULATIONS.  A  typical  circuit  for  the  56  tube  as  a  trans- 
former-coupled audio  amplifier  is  shown  in  Fig.  9  and  its  approximate  equivalent  in 
Fig.  10.  If  the  tube  used  in  Fig.  9  is  the  56  type  with  characteristics  as  shown  in  Fig.  8, 
and  its  various  constants  are  as  indicated  above,  the  constants  of  the  circuit  and  per- 
formance of  the  tube  may  be  calculated  as  follows: 

The  self  bias  resistor,  rlt  is  calculated  for  a  13.5-volt  operating  grid  bias  voltage  with 
5  ma  plate  current,  in  which  case,  according  to  the  curves  in  Fig.  8,  the  voltage  across 
the  terminals  marked  minus  and  plus  for  plate  supply  should  be  250  volts  and  the  actual 
plate  voltage  will  be  about  236.5.  The  value  of  the  resistor,  ri,  is  13.5/0.005  =  2700  ohms. 

The  by-pass  condenser,  Ci,  must  effectively  by-pass  n  at  the  lowest  audio  frequency 
desired  so  that  the  pulsating  plate  current  through  TI  will  not  generate  an  audio  voltage 
appreciable  compared  to  that  applied  to  the  grid.  The  voltage  across  rx  is  out  of  phase 
with  the  input  voltage  so  that  the  low-frequency  gain  may  be  lower  than  the  gain  at  other 


Type  56 


FIG.  9.     Transformer-coupled  Class  A  Amplifier 


FIG.    10.     Equivalent    Cir- 
cuit for  Fig.  9 


frequencies  if  the  impedance  of  C\  is  large.  The  capacitance,  Cz,  across  the  plate  sup- 
ply may  be  the  plate-supply  filter  condenser  but,  in  any  case,  must  have  sufficiently  low 
impedance  at  the  lowest  desired  frequency  to  by-pass  the  plate  supply  effectively  unless 
the  a-c  impedance  of  the  plate  supply  is  very  low. 

OUTPUT  CALCULATIONS.  The  dotted  resistance  r  in  Fig.  9  is  the  equivalent  im- 
pedance (as  nearly  unity  power  factor  as  possible)  of  the  primary  of  the  transformer, 
T,  with  the  load  resistance,  r&,  transferred  from  the  secondary  into  the  primary.  It  will 
be  noted  that  r  is  in  series  with  the  plate  of  the  tube  and  may  be  represented  by  a  load 
line  drawn  through  the  operating  point  at  5  ma  plate  current  and  13.5  volts  bias  in 
Fig.  8. 

The  proper  value  for  this  resistance  is  obtained  by  drawing  a  line  through  the  operating 
point  and  the  point  on  the  zero  grid  voltage  curve  representing  the  maximum  desired 
value  of  the  operating  plate  current  (in  this  case  9  ma,  since  experience  has  shown  that 
an  80  per  cent  change  in  plate  current  is  a  reasonable  value  to  assume).  A  check  along 
the  load  line  as  drawn  indicates  that,  if  the  grid  swings  in  a  negative  direction  to  double 
the  grid  voltage  of  13.5  volts,  the  decrease  in  plate  current  is  only  3.2  ma,  approximately. 


7-08 


VACUUM-TUBE   CIRCUIT  ELEMENTS 


This  load  line  represents  approximately  43,000  ohms  and  is  r  as  shown  in  Fig.  9.  The 
approximate  second  harmonic  under  4hese  conditions  may  be  calculated  by  the  following 
formula: 


H~  -^6  min  — 

- 


,  ,  ,  .  1   , 

per  cent  second  harmonic  approximately 

• 


-  -  -  -  —  =  pr  c  xi 

2(/  b  max  —  i~b  min)  ~  .-  ,   r      ^n  >  •  i    ,  >      (Q\ 

=  5.5  per  cent  for  80  per  cent  increase  in  plate  current  f      W 

as  assumed  above  J 

in  which  I&  is  the  "no-signal  plate  current,"  J&  max  is  the  peak  plate  current,  and  7$  min  is 
the  minimum  plate  current. 

Note  that  if  the  maximum  current  change  is  equal  on  each  side  of  the  operating  value 
the  second  harmonic  is  zero,  since  (I&  max  —  I&)  —  (1  &  —  /&  min)  =  0.  If  intermediate 
points  are  considered  and  it  is  found  that  equal  grid-voltage  increments  do  not  cause 
equal  plate-current  increments  throughout  the  range,  higher  harmonics  will  be  present. 
(See  p.  5-47.) 

The  average  power  output  may  be  calculated  by  the  following  formula,  which  is  readily 
derived  from  peak  values  of  a-c  voltages  and  currents. 


(•* 6  max  —  •*&  min)  (Eb  max  ~"  Eb  min) 
8 


power  output  or  0.28  watt  approximately,  I 
for  full  grid  swing  in  the  above  case          J 


in  which  Eb  max  is  the  maximum  plate  voltage  and  Eb  min  is  the  minimum  plate  voltage. 

The  theoretical  maximum  /&  max  for  the  class  A  amplifier  is  2I&,  and  the  minimum  J& 
is  zero.     The  corresponding  theoretical  maximum  Eb  max  is  2Eb,  and  the  minimum 
is  zero. 

The  plate  power  input  is  expressed  by  the  formula  Eblb  =  watts  input.     Therefore, 
the  plate  efficiency  at  maximum  output  is 


PL  eff.  = 
The  load  resistance  is 


b  max  —  ?b  min)  (Eb  max  ~  Eb  min) 


8EbIb 


=  23  per  cent 


Bin) 


(Ib  max  —  •*  b  min) 


43,000  ohms  approximately 


(ID 


(12) 


The  total  plate  voltage  swing  is  approximately  314  volts,  and  the  total  grid  swing  is 
27  volts,  which  is 

,-,  max ^  mn.  =  —=•  —  11.6  approximate  actual  amplification  by  the  tube     (13) 

\&c  max  —  &c  min)         ^' 

In  the  above  circuit  the  transformer  T  may  be  designed  as  a  voltage  coupling  trans- 
former to  supply  a  secondary  load  as  high  as  200,000  ohms  or  more,  an  output  impedance 
which  is  not  particularly  difficult  to  obtain  especially  if  a  choke  voltage  feed  is  used  as 
shown  in  Fig.  11,  so  that  the  d-c  plate  current  does  not  go  through  the  transformer  pri- 


FIG.  11.     Shunt  Transformer-coupled  Amplifier 

mary.  The  turn  ratio  would  be  the  square  root  of  the  impedance  ratio  or  a  turn  ratio 
of  the  primary  to  secondary  of  at  least  2.2.  The  total  amplification  of  this  amplifier 
stage  including  the  transformer  is  11.6  X  2.2  =  25.5,  approximately. 

The  above  condition  is  also  approximately  the  condition  for  maximum  power  output. 
If  the  calculated  distortion  is  too  high,  the  grid  swing  may  be  reduced  with  a  correspond- 
ing reduction  of  power  output.  In  actual  practice,  the  grid  swing  is  usually  very  small 
if  the  amplifier  is  used  as  a  voltage  amplifier  so  that  maximum  voltage  output  is  not  ob- 
tained but  the  voltage  amplification  is  the  same;  in  such  a  case  the  second  harmonic  may 
be  negligible  (far  less  than  1/2  per  cent) . 


CLASS  A  AMPLIFIERS  7-09 

PERFORMANCE  CALCULATIONS  FROM  TUBE  CONSTANTS.  Referring  to  the 
equivalent  circuit  shown  in  Fig.  10,  the  voltage  amplification  of  the  combination  of 
Fig.  9  and  Fig.  8  may  be  calculated  from  the  constants  of  the  tube  by  means  of  the  fol- 
lowing formulas: 


V.A. 


7/ 


+  r 

—  12.2  gain  without  transformer  (14) 

in  which  Eg  is  the  rms  grid  voltage  and  the  other  constants  are  as  denned  above. 

The  power  output  for  full  grid  swing  of  13.5  volts  peak  or  an  rms  value  of  9.6  volts  is 


P.  out  »     -     2  -  --  =  0.27  watt  (15) 

TP 

It  may  be  seen  that,  by  using  the  constants  of  the  tube  in  the  equivalent  circuit  of 
Fig.  10,  the  performance  calculations  do  not  differ  materially  from  the  performance  cal- 
culated from  the  curves  in  Fig,  8.  The  small  discrepancy  may  be  accounted  for  by  the 
fact  that  the  constants  of  ju  and  rp  used  in  eqs.  (14)  and  (15)  are  obtained  by  using  smaller 
increments  of  voltage  and  current  changes  than  are  used  in  eqs.  (10)  and  (13)  ;  in  other 
words,  the  effect  of  distortion  is  neglected. 

It  will  be  seen  that,  if  the  circuit  shown  in  Fig,  9  is  used,  the  only  difference  between  a 
voltage  and  power  amplifier  is  the  design  of  the  transformer,  T,  which,  in  a  voltage  ampli- 
fier works  into  a  high  resistance  and  in  a  power  amplifier  works  into  the  desired  load, 
the  turn  ratio  being  chosen  to  present  the  same  equivalent  primary  impedance. 

If  in  the  resistance  coupled  amplifier  the  load  resistance  is  43,000  ohms,  the  gain  is 
the  same  as  calculated  for  the  above  case  if  TK  ^>  r.  The  same  load  line  as  shown  in  Fig.  8 
may  be  used  provided  the  plate  supply  voltage  is  equal  to  the  value  indicated  by  the  in- 
tersection of  the  load  line  and  the  zero  plate-current  axis.  This  plate  voltage  is  approxi- 
mately 460  volts  instead  of  the  250-volt  supply  for  the  transformer-coupled  case. 

POWER  OUTPUT  AND  PLATE  EFFICIENCY.  In  the  operation  of  vacuum  tubes 
as  amplifiers  (also  as  oscillators,  or  detectors)  at  low  power  levels  it  is  usually  desirable 
to  get  as  much  output  signal  power  as  possible  from  a  given  tube  regardless  of  the  plate 
efficiency  (the  ratio  of  the  output  signal  power  to  the  input  plate  supply  power).  Under 
such  conditions  there  is  usually  one  of  two  parameters  limiting  the  power  output,  namely, 
the  input  signal  voltage  in  amplifiers  and  detectors,  or  the  plate  supply  potential  in  the 
case  of  all  vacuum-tube  operation. 

When  the  input  signal  voltage  is  the  limiting  factor,  and  the  tube  is  operating  over  a 
linear  part  of  its  characteristic  (for  detectors  this  means  linear  with  respect  to  its  detec- 
tion characteristic  —  see  Detectors,  p.  7-76)  ,  the  vacuum  tube  can  be  considered  a  gener- 
ator with  an  internal  resistance.  Then  the  maximum  power  output  occurs  when  the 
load  is  a  pure  resistance  equal  to  the  internal  resistance. 

Frequently  the  input  voltage  available  is  large  enough  so  that  for  the  given  plate  sup- 
ply voltage  the  introduction  of  distortion  is  the  limiting  factor.  Using  the  Taylor's 
series  development  of  the  plate  current  up  to  and  including  the  second  power  term,  War- 
ner and  Loughren  (Proc.  I.R.E.,  Vol.  13,  709  [1925])  showed  that  under  these  conditions 
the  maximum  undistorted  power  output  is  obtained  when  the  load  was  resistive  and 
equal  to  twice  the  internal  resistance.  Experiment  has  generally  shown  this  to  be  ap- 
proximately correct  for  amplifiers  using  low  amplification  triodes  in  class  A  operation 
but  that  no  such  simple  rule  applies  in  high-mu  triodes,  tetrodes,  pentodes,  and  class  B 
operation  of  triodes.  The  optimum  value  of  load  resistance  is  usually  determined  ex- 
perimentally and  forms  part  of  the  operating  data  furnished  by  the  tube  manufacturer. 

When  vacuum  tubes  are  operated  at  high  power  levels  it  is  usually  economically  neces- 
sary to  consider  the  plate  efficiency.  This  is  given  by  the  formula 


Eff. 


where  to  is  the  component  of  the  current  in  the  plate  circuit  having  the  desired  output 
frequencies  and  amplitudes;  eo  is  the  corresponding  voltage  drop  across  the  load  ZL,  the 
resistance  component  of  which  is  TL;  t  is  the  instantaneous  value  of  the  plate  current; 
JEb  is  the  plate  supply  potential;  and  T7  is  a  complete  period  of  i. 


7-10 


VACUUM-TUBE   CIRCUIT  ELEMENTS 


When  the  current  i  is  given  in  the  form 

i  =  IQ  +  /i  cos  cat  +  /2  cos  2cat  •+• 
the  plate  efficiency  becomes 

Eff.  -^ 


FIG.  12.     Pushpull  Amplifier 


when  the  fundamental  is  the  desired  frequency  of  the  output  power. 

PUSHPULL  AMPLIFIER.     Figure  12  represents  a  pushpull  amplifier,  which  has  sev- 
eral advantages  over  the  single-ended  amplifier.     A  class  A  pushpull  amplifier  does  not 

take  a  pulsating  current 
from  the  plate  supply  for 
the  ideal  condition,  and  in 
practice  the  variation  of 
supply  plate  current  is  neg- 
ligible. This  essentially 
constant  plate  current  per- 
mits the  omission  of  the  by- 
pass condenser  that  is  re- 
quired in  a  single-ended 
class  A  amplifier,  shown  as 
Ci  in  Fig.  9.  Another  ad- 
vantage of  the  pushpull  am- 
plifier is  that  the  d-c  magnetizing  current  in  the  primary  of  the  plate  transformer  is 
balanced  out,  which  simplifies  the  plate  transformer  design.  (See  p.  6-17.) 

The  equivalent  circuit  of  the  pushpull  class  A  amplifier  is  shown  in  Fig.  13,  which  is 
similar  to  the  circuit  shown  in  Fig.  10,  except  that  the  resistances  of  both  plates  are  in 
series  with  the  load  resistance.  In  order  that  the  load  resistance  r  of  43,000  ohms  be 
obtained  for  the  56-type  tube  as  discussed  for  the  single-ended  amplifier,  it  is  necessary 
that  the  primary  impedance  of  the  loaded  output  transformer  be  86,000  ohms,  which  is 
not  very  practical  for  voltage  amplifier  purposes,  because  of  the  large  number  of  turns 
required  in  the  primary.  Another  disadvantage  of  the  pushpull  voltage  amplifier  is  that 
one-half  of  the  total  input  voltage  is  available  for  each  grid  and  the  increased  power  out- 
put increases  the  output  voltage  into  a  given  resistance 
by  only  approximately  40  per  cent.  Therefore,  the 
actual  decrease  in  voltage  gain  by  using  two  tubes  in 
pushpull-instead  of  one  single-ended  tube  is  about  20  per 
cent. 

The  calculations  of  power  output  of  the  class  A  push- 
pull  amplifier  may  be  made  by  using  eq.  (10)  or  (15),  if 
the  results  are  multiplied  by  2.  The  load  resistance  is 


\ 


•-wwvv- 


FIG.    13.     Equivalent    Circuit    for 
Class  A  Pushpull  Amplifier 


calculated  by  multiplying  eq.  (12)  by  2  so  that  the  transformer  primary  impedance  (plate 
to  plate)  is  double  the  value  obtained  for  a  single-ended  amplifier. 

TRIODE  VOLTAGE  AMPLIFIER.  The  class  A  voltage  amplifier  in  general  uses  a 
special  high-gain  type  of  tube  instead  of  the  general-purpose  tube  such  as  the  56.  The 
tube  characteristic  desired  is  indicated  by  eqs.  (13)  and  (14) .  The  higher  the  amplifica- 
tion constant  ju,  other  constants  being  equal,  the  higher  will  be  the  voltage  amplification. 
Tubes  designed  primarily  for  voltage  amplifiers  of  the  triode  type  are  the  240,  841,  85, 
203A,  and  others.  The  general-purpose  tube  such  as  the  56  discussed  in  some  detail 
also  has  quite  extensive  application  as  a  voltage  amplifier,  where  interstage  coupling 
transformers  are  used.  In  general,  the  plate  resistance  of  the  high-amplification-factor 
tubes  is  quite  high,  so  that  such  tubes  are  employed  principally  in  resistance-coupled  ampli- 
fiers. Because  of  the  general  use  of  screen-grid  tubes  as  high-gain  amplifiers  the  triode  is 
becoming  less  common  as  a  voltage  amplifier  except  where  coupling  transformers  are  used. 

TRIODE  POWER  AMPLIFIER.  The  above  discussion  applied  primarily  to  the 
triode  as  a  class  A  amplifier  in  which,  in  general,  the  grid  is  not  driven  into  the  positive 
region.  With  this  limitation  for  grid  excitation,  it  is  seen  from  eq.  (10)  that  the  higher 
the  plate-current  swing  for  a  given  plate-voltage  swing  the  greater  will  be  the  power  out- 
put and  correspondingly  the  greater  the  plate  dissipation.  Therefore,  a  power  tube  to 
operate  as  a  class  A  amplifier,  without  positive  grid  swing,  should  be  a  low-plate-resistance 
tube  with  a  correspondingly  low  amplification  constant  and  capable  of  relatively  high 
plate  power  dissipation.  The  various  types  of  triode  tubes  primarily  designed  for  the- 
class  A  power  output  service  are  the  31,  45,  2 A3,  842,  250,  845,  849,  and  848.  The  type- 
31  tube  is  the  small  battery  tube  rated  as  0.375  watt  output;  the  848  is  a  large  water- 
cooled  tube  rated  at  approximately  1900  watts  output  as  a  class  A  amplifier. 


CLASS  A  AMPLIFEBES 


7-11 


200        300         400 
Plate  Votts 

FIG.  14.     Type  2A3  Plate  Characteristics 


A  typical  low-plate-resistance  tube  designed  as  a  class  A  power  output  tube  is  the  2A3, 
the  plate  characteristics  of  which  are  shown  in  Fig.  14.  Calculations  similar  to  those 
made  on  the  56  tube  indicate  that  about  3.5  watts  output  can  be  obtained  from  the  2A3 
with  a  load  resistance  of  2500  ohms,  a  bias  of  43.5  volts,  a  plate  voltage  of  250,  and  a 
plate  dissipation  of  15  watts  at  zero  out- 
put. The  plate  dissipation  decreases  from 
full  value  at  no  signal  to  the  full  value 
less  the  output  power  with  signal.  The 
above  15-watt  plate  dissipation  for  the 
2 A3  decreases  to  11.5  watts  at  full  output. 
PENTODE  VOLTAGE  AMPLIFIER. 
The  class  A  pentode  amplifier  may  be 
used  as  a  voltage  amplifier  for  audio  or 
radio  frequencies.  The  fundamental  cal- 
culations of  power  output  and  determina- 
tion of  load  (r)  lines  is  essentially  as  dis- 
cussed for  the  triode  amplifier,  but  certain 
details  of  the  calculations  are  different. 
One  of  the  principal  differences  is  that 
the  amplification  factor  is  much  higher 
than  the  amplification  obtained  in  prac- 
tice because  the  plate  resistance,  rp,  of 
the  tube  is  much  higher  than  it  is  feasible 
to  equal  with  a  load  resistance,  r.  Because  of  the  very  high  amplification  factor  of  pen- 
todes as  well  as  tetrodes  with  a  correspondingly  high  plate  resistance,  eqs.  (10),  (12), 
and  (13)  are  much  more  useful  than  eq.  (14).  The  transconductance,  gm,  for  the  pen- 
todes and  tetrodes  is  also  quite  useful  in  performance  calculations. 

Referring  to  the  definition  and  means  by  which  the  transconductance  was  obtained 
(as  given  in  Section  4) ,  it  will  be  noted  that  the  condition  for  obtaining  data  for  gm  cal- 
culations is  that  the  resistance  in  series  with  the  plate  is  zero  or  at  least  sufficiently  low 
not  to  alter  the  observed  plate-current  change  for  a  given  grid-voltage  change.  It  may 
be  seen  from  the  plate  current  characteristic  of  a  type-57  pentode,  as  shown  in  Fig.  15, 
that  the  slope  of  a  load  resistance  line  drawn  through  the  indicated  operating  point  does 
not  alter  the  plate-current  change  appreciably  for  a  given  grid  voltage  swing.  This  tube 
is  used  extensively  for  radio-frequency  and  audio-frequency  high-gain  amplifiers.  Ex- 
cept where  maximum  power  or  voltage  swing  is  desired,  it  is  not  necessary  to  use  the 

plate-current  curves  to  make 
performance  calculations  (ex- 
cept to  prevent  overloading), 
as  such  calculations  can  be 
readily  made  from  the  con- 
stants of  the  tube.  For  the 
conditions  as  given  in  Fig.  15, 
the  transconductance  at  —3 
volts  bias  is  1225  micromhos, 
the  plate  resistance  is  given 
as  greater  than  1.5  megohms, 
and  the  amplification  factor 
is  greater  than  1500.  It  is 
obvious  that,  with  the  more 
or  less  indefinite  amplification 
factor  and  plate  resistance, 
the  equations  involving  these 
constants  are  of  little  value  in 
calculating  the  voltage  gain 
in  a  57  amplifier.  However, 


100 


200  300 

Plate  Volts 


400 


Screen  VoTts  »  100  Suppressor  Volts  »  2 

FIG.  15.     Type  57,  Plate  Characteristics 


since  the  gm  of  the  tube  is  definite  within  limits  and  the  above  constants  are  high,  the 
plate-current  change  for  a  given  signal  EK  is:  Eggm  —  I?  and  I^r  —  Ev.  Therefore,  the 
voltage  gain  is 

-£  =  gmr  =  Voltage  gain  (16) 


For  the  assumed  conditions  and  a  load  of  100,000  ohms  the  voltage  gain  is 
1225  X  10"6  X  100,000  =  122.5 


7-12 


VACUUM-TUBE   CIRCUIT  ELEMENTS 


The  above  value  of  voltage  amplification  is  approximately  6  per  cent  high  if  the  plate 
resistance  of  the  tube  is  1.5  megohms,  but  such  corrections  are  usually  unnecessary  unless 
great  accuracy  is  required,  and  then  it  is  advisable  to  measure  the  transconductance  of 
the  tube,  as  well  as  the  plate  resistance,  at  the  point  of  actual  operation.  A  decrease  of 
load  resistance  to  50,000  ohms  reduces  the  gain  to  approximately  50  per  cent  of  the  above 
calculated  value,  and  the  accuracy  is  greater  because  of  the  lower  value  of  load  resistance 
relative  to  the  actual  plate  resistance. 

In  an  audio  amplifier  it  is  possible  to  attenuate  the  high  audio  frequencies  considerably 
if  the  load  or  coupling  resistance  is  sufficiently  high  so  that  the  shunting  capacitances 

appreciably  lower  the  imped- 
ance of  r.  These  capacitances 
are  the  plate  to  suppressor  grid 
capacitance  of  the  57  amplifier 
plus  the  capacitance  of  the  tube 
or  device  to  which  the  amplifier 
is  coupled.  The  output  capaci- 
tance of  a  57-type  tube  as  an 
audio  amplifier  is  about  6.5  y.yJL , 
and  the  input  capacitance  to  a 
similar  following  stage  of  am- 
plification is  about  5  wi  or  a 
total  of  about  12  /j/zf,  the  im- 
pedance of  which  is  about  1.3 
megohms  at  10,000  cycles, 
which  would  not  appreciably 
affect  the  above  57  amplifier 
with  an  r  of  100,000  ohms,  at 
the  higher  audio  frequencies. 


•300 


FIG.  16.     Type  58,  Plate  Characteristics 


1500 


Plate 


Volts  «250 


Screen  Volts=100 

Suppressor  Volts=rO 


As  commonly  used  the  voltage  output  from  a  57  amplifier  is  so  small  that  little  non-linear 
distortion  is  introduced. 

If  the  tube  is  used  as  a  radio-frequency  amplifier,  the  plate  impedance  is  obtained  by  a 
parallel-resonant,  or  an  equivalent,  circuit  in  series  with  the  plate,  and  for  known  equiva- 
lent series  impedance  of  the  plate  circuit  the  voltage  gain  may  be  calculated  as  indicated 
above.  Since  the  tube  resistance  is  very  high,  the  selectivity  of  the  amplifier  for  tuned 
output  is  essentially  the  selectivity  of  the  tuned  plate  circuit. 

VARIABLE-GAIN  PENTODE  VOLTAGE  AMPLIFIER.  The  57  tube  just  discussed 
is  not  well  adapted  to  use  as  a  variable-gain  control  tube  by  changes  in  the  bias.  A  tube 
designed  primarily  for  a  bias-controlled  vari- 
able-gain amplifier  is  known  as  the  type-58 
tube,  the  characteristics  of  which  are  shown 
in  Figs.  16  and  17.  Since  the  plate  resistance 
of  the  tube  is  approximately  800,000  ohms  or 
more,  the  gain  as  calculated  by  eq.  (16)  is  quite 
accurate  and  the  gain  decreases  as  the  negative 
grid-bias  increases  because  the  transconduct- 
ance decreases  for  increased  bias.  The  distor- 
tion in  the  output  of  this  tube  is  greater  than 
for  the  57-type  tube,  and  for  this  reason  the 
58-typeisnot  generally  used  for  an  audio  voltage 
amplifier.  If  the  plate  circuit  of  the  58  is  tuned 
and  the  grid-voltage  swing  is  not  too  great,  little 
distortion  results  from  the  use  of  this  tube  in 
the  radio-frequency  system  of  a  receiver.  The 
bias  may  be  supplied  from  an  automatic  vol- 
ume-control system,  the  output  of  which  supplies  Fm>  17  T  58>  Transconductance 
a  bias  that  is  proportional  to  the  carrier 

value  of  the  received  signal,  or  may  be  controlled  manually,  or  by  a  combination  of  the 
two.  The  type-58  or  other  tubes  with  the  approximate  exponential  grid  voltage  vs.  plate 
current  characteristics  (such  as  types  34,  35,  39,  78,  6C6)  are  almost  universally  employed 
in  receivers  so  that  a  simple  bias  control  may  be  used  to  control  the  sensitivity  of  the 
radio-frequency  amplifier.  (See  Ballantine  and  Snow,  Proc.  I.R.E.,  Vol.  18,  No.  12, 
p.  2102  [December,  1930].)  Such  a  system  permits  an  automatic  volume  control  to  de- 
termine the  output  level  to  the  audio  system  so  that  all  stations  above  a  predetermined 
level  and  with  a  given  percentage  of  modulation  will  be  reproduced  at  essentially  the 
same  volume. 


21000 


500 


40       30       20 
Control  Grid  Volts 


CLASS  A  AMPLIFIERS 


7-13 


It  will  be  noted  that,  of  the  tubes  listed,  all  are  pentodes  except  the  early  exponential 
tube,  type  35.  The  principal  reason  for  the  pentode  construction  is  that  the  plate  voltage 
swing  is  not  limited  by  the 
screen-grid  voltage  for  the 
pentode  as  for  the  screen-grid 
tube. 

PENTODE  POWER  AM- 
PLIFIER. This  latter  charac- 
teristic of  pentode  tubes  per- 
mits comparatively  large  power 
outputs  from  tubes  designed 
primarily  as  audio  power  out- 
put tubes.  The  power  pentode 
tubes  are  used  principally  as 
audio  power  amplifiers  in  radio 
receivers;  however,  the  power- 
type  pentode  designed  for  use 
in  transmitters  is  gaining 
favor,  but  not  for  class  A  am- 
plifiers. 


480 


FIG.  18.     Type  47,  Plate  Characteristics 


The  first  power  pentode  tube  designed  for  the  output  system  of  receivers  was  the  type 
47,  but  it  is  being  replaced  by  the  indirectly  heated  cathode  pentodes  such  as  types  2A5, 
42,  and  41.  The  characteristics  of  the  47,  however,  are  typical  of  the  power  pentodes. 
The  plate  characteristics  for  the  47  are  shown  in  Fig.  18,  and  power-output  calculations 
for  full  grid  swing  may  be  made  by  using  eq.  (10).  The  load  line  drawn  through  the 
operating  bias  of  15.3  volts  at  250  volts  on  the  plate  and  screen  grid  represents  7000  ohms 
and  is  approximately  the  load  resistance  for  maximum  power  output  for  full  grid  swing 

and  mmjm^rrn  distortion.  A  typ- 
ical circuit  for  the  pentode  class 
A  power  amplifier  is  shown  in  Fig. 
19,  for  the  47-type  tube.  The  cal- 
culated power  output  according  to 
eq.  (10)  is  approximately  2.4  watts. 
Two  tubes  may  be  used  in  push- 
pull  similar  to  Fig.  12.  The  de- 
sirable characteristic  of  the  pentode 
tube  is  its  power  sensitivity,  that  is, 
the  ratio  of  power  output  per  volt 


FIG.  19.     Circuit  for  Single  Type-47  Audio  Amplifier 


of  signal  applied  to  the  grid.     This  ratio  is  3  to  4  times  as  large  as  for  triodes,  but  the  pentode 
power  tube  has°several  peculiar  characteristics  which  are  not  particularly  desirable. 

Distortion  in.  Pentode  Audio  Power  Amplifier.  Referring  to  Fig.  18,  it  will  be  noted 
that  the  plate-current  increase  for  a  peak  grid  swing  of  15.3  volts  (the  value  of  the  bias) 
is  practically  equal  to  the  plate-current  decrease  for  the  same  grid-voltage  swing  in  the 
negative  direction  along  the  7000-ohm  load  line.  Therefore,  the  second  harmonic  is  es- 
sentially zero,  but  it  can  also  be  seen  that  the  plate-current  change  per  volt  change>tin 
bias  near  the  extreme  swings  of  the  bias 
decreases,  which  produces  a  flat-topped 
wave.  The  predominating  harmonic  in 
such  a  wave  is  the  third,  but  other  har- 
monics may  be  present,  their  value  de- 
pending on  the  flatness  of  the  top  of  the 
output  wave.  (See  p.  5-47.)  It  is  usu- 
ally much  simpler  and  more  accurate  to 
measure  the  harmonic  content  under  ac- 
tual operating  conditions  than  it  is  to  cal- 
culate it.  All  the  unknown  factors,  such 
as  inefficiency  of  output  transformer 
and  lack  of  proper  plate  supply  filtering, 
are  accounted  for  in  the  measured  val- 
ues; their  effects  are  very  difficult  to  de-  ' 
termine  and  express  mathematically. 
Experimental  curves  of  percentage  dis- 
tortion of  a  47  tube  are  shown  in  Fig.  20. 

BIDIRECTIONAL  AMPLIFIERS—REPEATERS.    All  the  circuits  considered  above 
amplify  in  one  direction  only  and  will  not  pass  appreciable  energy  in  the  other  direction. 


Power  Output-  Walls 
_O  1-1  ro  Q 

<& 

^ 

^ 

\r 

I 

< 

•y£~ 

~~i 

I 

V 

1 

Grid  V 
Screen 
Plate  X 
Grid  S3 
Volte 

:rts—  16*5 
Vofts=250  . 
folts=25Q 
snaF=15-3 
Peak 

s 

'\ 

k 

V 

3                  400O              8000             220OO 
Load  Resistance,  rp<,  Obese 

FIG.  20.     Type  47,  Output  Characteristics 


7-14 


VACUUM-TUBE   CIRCUIT  ELEMENTS 


West 


There  are  some  applications,  notably  two-way  telephony  over  wires,  where  it  is  desirable 
to  amplify  in  either  direction.     (See  also  Section  17,  p.  39.) 

Figure  21  shows  a  schematic  diagram  of  a  circuit  which  amplifies  signals  coming  from 
either  end  of  the  circuit,  line  West  or  East,  and  distributes  the  amplified  signals  equally 
between  the  two  lines.  Inclosed  within  the  dotted  line  is  a  three-winding  transformer; 
as  shown  in  article  7,  p.  6-12,  if  the  proper  circuit  adjustments  are  made,  energy  coming 
from  either  line  is  divided  by  the  three-winding  transformer  equally  between  the  input 

circuit  (connected  across 
A-B}  and  the  output  cir- 
cuit (inductively  con- 
nected). The  half  enter- 
ing the  output  circuit  is 
dissipated  as  heat;  but  the 
half  entering  the  input  cir- 
cuit is  introduced  into  the 
grid  circuit  of  a  vacuum 
tube  through  a  step-up 
transformer  and  a  poten- 
tiometer. This  is  ampli- 
fied by  the  tube,  or  tubes, 
and  appears  in  the  out- 
put winding  of  the  three- 
winding  transformer 
where  it  is  again  divided 
equally  between  East  and 


FIG.  21.     Bidirectional  Amplifier 


West  lines.  Thus,  although  the  incoming  energy  is  twice  divided,  so  that  only  one- 
fourth  of  it  is  utilized,  the  power  amplification  of  the  tube,  or  tubes,  may  easily  be  several 
hundred,  so  that  the  net  result  is  a  great  increase  in  energy  over  the  original. 

The  chief  practical  difficulty  with  this  circuit  is  that  the  impedance  of  lines  West  and 
East  must  be  identical  at  all  frequencies,  or  energy  will  be  fed  by  the  three-winding  trans- 
former from  the  output  to  the  input  circuit  and  the  tube  will  furnish  sustained  oscilla- 
tions, or  "sing." 

To  avoid  the  requirement  of  working  between  similar  impedances  a  circuit  such  as  is 
shown  in  Fig.  22  is  used.  Here  a  network  of  resistors,  inductors,  and  condensers  is  de- 
signed to  balance  the  external  impedance  on  each  side  of  the  amplifier.  The  principle 
of  operation  is  the  same  as  above,  the  left-hand  tube  amplifying  signals  entering  from  the 
line  West  and  dividing  the  energy  between  line  East  and  the  accompanying  network  (JV) . 


Output 


Output 


Fro.  22.     Bidirectional  Amplifier  with  Balancing  Networks 

Similarly,  signals  entering  from  line  East  are  amplified  by  the  right-hand  tube  and  divided 
between  line  West  and  its  network.  An  impedance  unbalance  between  either  line  and  its 
associated  network  causes  some  of  the  energy  of  the  output  circuit  to  be  fed  by  the  three- 
winding  transformer  into  its  input  circuit,  where  it  is  amplified  and  fed  into  the  other 
transformer.  An  unbalance  here  will  similarly  cause  some  of  the  energy  to  find  its  way 
back  through  the  original  path.  Thus  it  is  seen  that  if  both  lines  are  poorly  matched 
by  their  networks  the  amplifier  may  act  as  an  oscillator,  but  if  either  line  is  perfectly 
balanced  by  its  network  no  sustained  oscillations  will  be  produced. 


CLASS  B  AMPLIFIERS  7-15 

Amplifier  circuits  of  this  type  are  in  extensive  use  by  telephone  and  telegraph  com- 
panies on  their  long  lines.  They  are  called  repeaters, 

SUMMARY  FOR  CLASS  A  AMPLIFIERS.  The  foregoing  discussion  of  the  class  A 
amplifiers  and  the  various  types  of  tubes  for  particular  services  assumed  that  the  grid 
swing  or  excursions  were  always  in  the  negative  region,  and  that  the  amplifier  behaved 
in  general  according  to  the  definition  of  the  class  A  amplifier,  as  given  at  the  beginning 
of  this  section  on  amplifiers.  There  are  a  few  facts  about  the  characteristics  of  class  A 
amplifiers  which  should  be  kept  in  mind  when  designing  or  studying  such  devices;  some 
of  them  are  listed  below. 

1.  The  maximum  voltage  output  of  a  class  A  amplifier  measured  at  the  plate  of  the 
tube  is  limited  by  the  plate-voltage  limitations  of  the  tube  and  the  plate  load,  r. 

2.  The  maximum  power  output  from  a  class  A  amplifier  is  limited  by  the  plate  dissipa- 
tion at  zero  output  and  the  TninJTrmm  instantaneous  plate  voltage  for  peak  positive  grid 
swing  to  zero,  at  which  the  plate  current  is  approximately  double  the  steady  plate-current 
value. 

3.  The  load  impedance  into  which  the  tube  must  work  for  maximum  power  output 
is  not  a  simple  function  of  the  plate  resistance  of  the  tube  but  depends  upon  the  two 
conditions  above  in  cases  where  the  grid  swing  is  not  limited.    Power-output  tubes  have 
high  bias  and  low  plate  resistance  so  that  grid  swings  to  zero  voltage  are  not  the  controlling 
limitation. 

4.  The  power  output  for  a  comparatively  low  and  limited  grid  swing,  that  is,  when 
limitations  of  1  and  2  are  not  effective,  is  maximum  when  the  load  resistance  is  equal  to 
the  plate  resistance  of  the  tube.    For  this  same  grid-swing  limitation,  the  maximum  out- 
put voltage  or  gam  approaches  the  amplification  factor  of  the  tube  as  the  load  resistance, 
r,  increases  toward  the  value  of  the  tube  plate  resistance,  rp. 

2.  CLASS  B  AMPLIFIERS 

GENERAL.  Because  of  the  relatively  low  power  output  of  class  A  amplifiers  and  the 
fact  that  the  plate  dissipation  is  maximum  for  no-signal  conditions,  the  overall  efficiency 
of  such  an  amplifier  is  very  low.  In  applications  where  considerable  power  is  desired  either 
at  radio  or  audio  frequencies,  the  size  of  the  tubes  and  the  cost  of  plate  power  supply,  per 
watt  output  for  class  A  amplifiers,  increases  so  rapidly  that  such  amplifiers  are  not  used. 
In  the  broadcast  receiver  it  is  not  economical  to  obtain  power  outputs  greater  than  about 
5  or  6  watts  without  resorting  to  class  B  amplifiers  or  some  combination  of  the  class  A 
and  B  amplifiers. 

The  grid  swing  in  the  class  A  amplifier  is  usually  limited  to  the  negative  region  for  the 
entire  input  cycle  because,  in  general,  grid  swings  into  the  positive  region  result  hi  plate- 
current  distortion,  chiefly  because  of  the  large  external  grid  circuit  impedance  and  limita- 
tions in  plate-current  swing. 

According  to  the  definition  of  a  class  B  amplifier,  the  bias  is  such  that  the  operating 
plate  current  is  small,  so  that  for  the  no-signal  condition  the  plate  dissipation  is  low. 
Therefore,  the  grid  swing  in  general  is  limited  only  by  a  non-linearity  of  plate  current 
and  grid  voltage  when  the  grid  swings  positive  from  the  operating  point.  Almost  any 
three-element  tube  and  some  pentodes  may  be  used  in  a  class  B  amplifier.  However, 
some  tubes  are  to  be  preferred  for  reasons  that  will  be  evident  after  the  various  calcula- 
tions are  made  from  the  various  sample  tube  characteristics  as  shown  and  discussed 
below. 

LOW-POWER  AUDIO  AMPLIFIERS.  Since  self-bias,  as  obtained  in  a  pushpull 
circuit  as  shown  in  Fig.  12,  depends  upon  the  signal  (because  the  average  plate  current  in  a 
class  B  amplifier  depends  upon  the  signal)  and  since  there  is  no  other  convenient  means  for 
supplying  fixed  bias  in  the  usual  receiver,  low  current  at  zero  bias  is  very  desirable  in  tubes 
for  use  in  class  B  amplifiers.  The  plate  characteristic  curves  for  such  a  tube  are  shown 
in  Fig.  23,  with  the  corresponding  grid  currents  for  various  grid  voltages  plotted  against 
plate  voltage.  Experience  has  indicated  that  a  plate  load  resistance  of  about  2000  ohms 
for  this  tube  with  400  volts  on  the  plate  is  approximately  the  optimum  value  for  power 
output  and  plate-circuit  efficiency.  The  2000-ohm  load  line  is  drawn  through  the  operat- 
ing point  to  obtain  data  for  power-output  calculations  and  grid-current  values  to  deter- 
mine the  input  resistance  of  the  grids.  Data  can  be  obtained  from  this  set  of  curves  to 
replot  curves  of  plate  current  and  grid  current  against  grid  voltage,  which  are  more  useful 
for  performance  calculations  than  the  curves  in  Fig.  23.  Such  curves  are  commonly 
called  dynamic  transfer  characteristics. 

The  data  for  these  dynamic  transfer  characteristics  may  be  optionally  taken  directly 
from  meter  readings  in  a  circuit  such  as  shown  in  Fig.  24.  Here  are  plotted  various  curves 


7-16 


VACUUM-TUBE   CIRCUIT  ELEMENTS 


for  different  load  resistances.  Except  at  points  near  the  zero  bias  axes,  that  is,  for  a  grid 
swing  of  about  5  volts,  these  curves  represent  the  dynamic  performance  of  the  46  tube  for 
various  load  resistances  for  one  half-cycle.  Another,  similar  tube  is  connected  in  pushpull 


'  200  i 


360      400      44Q 


FIG.  23.     Type  46,  Plate  Characteristics 


in  order  to  supply  power  for  the  other  half-cycle.  If  the  two  tubes  have  similar  character- 
istics and  the  pushpull  input  and  output  transformers  are  well  balanced,  the  output  wave 
will  be  symmetrical  and  only  odd  harmonics  will  be  present  in  the  output  for  a  sinusoidal 
input  wave. 

The  power-output  calculations  for  the  46  from  the  curves  in  Fig.  24  are  made  like  the 
power-output  calculation  for  the  class  A  amplifier,  except  that  the  plate  current  Imax 
(the  peak  plate  current  read  from  the  curves)  is  the  total  change  in  plate  current  from  the 
operating  point  of  essentially  zero  plate  current.  Since  Jmax  is  the  peak  value  of  the  out- 
put wave,  and  if  little  distortion  is 
present  for  sinusoidal  inputs,  the  power 
output  for  the  two  tubes  is  72max^/2. 
The  average  plate-current  input  for 
two  tubes  at  a  constant  plate  supply 
voltage,  JSb,  for  approximately  full  power 
output  is  /maxS/T  =  0.637Imax-  Since 
the  plate  voltage  is  constant  the  power 
input  to  the  plates  of  the  two  tubes  is 
0.637Imax  JEb.  Therefore,  the  plate  effi- 
ciency is 

"•"•-rig      <»? 

and  the  plate  dissipation  per  tube  is 


PL  loss  = 


-  0.5/2maxr 


(17a) 


-10 


+10 


H-20     4-30    +40 
Grid  Volts 


+50    +60     +70 


FIG.  24.    Type  46,  Dynamic  Transfer  Characteristics 


The  maximum  theoretical  efficiency 
is  obtained  when  the  peak  a-c  plate 
voltage  is  equal  to  the  d-c  plate  volt- 
age and,  according  to  eq.  (17),  is  78.5 
per  cent. 

The  above  equations  are  quite  accu- 
rate at  full  or  nearly  full  grid  swings; 


the  accuracy  decreases  with  lower  grid  swings,  the  decrease  in  accuracy  being  a  function 
of  the  no-signal  or  standby  plate  current.  For  the  theoretical  condition  for  a  class  B 
amplifier  of  zero  plate  current  at  no  signal,  they  are  accurate  for  all  grid  swings. 


CLASS  B  AMPLIFIERS 


7-17 


If  the  maximum  value  of  plate  current  is  taken  from  the  curves  in  Fig.  24,  for  the 
2000-ohm  load,  sample  calculations  are  as  follows: 


0.1502  X  2000 


=  22.5  watts  output 


0.637  X  0.150  X  400  - 


0.150  X  2000 
1.27  X  400 

0.1502  X  2000 


59  per  cent  plate  efficiency 


7.9  watts  plate  loss  per  tube 


It  should  be  noted  that  the  load  resistance  used  is  the  resistance  the  tube  works  into 
during  the  half-cycle  it  functions  so  that  the  plate  to  plate  equivalent  load  on  the  primary 
of  the  output  transformer  is  4r. 

INPUT  RESISTANCE.  The  non-linear  grid  current  in  class  B  amplifiers  for  audio 
frequencies  is  a  principal  difficulty  to  overcome.  The  plate-current  grid-voltage  curves, 
as  shown  in  Fig.  24,  are  for  perfect  regulation  of  grid  voltage  or  the  equivalent  of  zero 
resistance  in  series  with  the  grids.  An  essentially  zero  input  resistance  to  the  grids, 
however,  is  not  practical  to  attain,  but  values  can  be  obtained  which  are  quite  low  com- 
pared to  the  minimum  instantaneous  resistance  of  the  grids  of  the  class  B  amplifier  tubes. 

The  slope  of  the  grid-current  curves  in  Fig.  24  at  any  particular  grid  voltage  gives  the 
instantaneous  input  resistance  of  the  amplifier.  The  niinimum  value  of  the  grid  resistance 
occurs  at  a  maximum  grid  swing  of  about  45  volts  positive.  The  slope  of  the  grid-current 
curve  at  this  point  represents  approximately  500  ohms,  while  at  points  below  45  volts 
on  the  grid,  the  grid  resistance  is  about  1600  ohms,  and  at  near  the  zero  axis,  the  resistance 
is  still  higher  for  each  tube.  However,  the  grid  current  is  not  zero  at  zero  bias,  so  that 
the  input  resistance  of  the  two  tubes  over  the  region  at  which  grid  current  flows  to  each 
tube  is  the  resistance  of  the  two  tubes  in  parallel.  Therefore,  to  drive  the  grids  of  two 
46-type  tubes  properly  as  audio  amplifiers,  the  amplifier  stage  supplying  voltage  to  the 
grids  of  the  46's  works  into  a  load  resistance  of  approximately  4500  ohms  for  each  tube, 
or  a  combined  resistance  of  approximately  2250  ohms  for  a  short  period  during  which 
each  tube  draws  grid  current.  When  one  tube  ceases  to  draw  grid  current,  because  its 
grid  is  too  negative,  the  input  resistance  rises  very  rapidly  to  approximately  4500  ohms, 
after  which  it  decreases  gradually  to  500  ohms. 

Because  of  this  erratic  change  of  input  resistance  to  the  46's  as  class  B  audio  amplifiers, 
it  is  very  important  to  keep  the  equivalent  impedance  in  series  with  each  grid  a  minimum. 
The  low-impedance  requirement  is  met  by  using  low-plate-resistance  driver  tubes,  prefer- 
ably in  pushpull  with  a  transformer  step-dawn  ratio  as  great  as  the  plate-voltage  swing 
of  the  driver  tubes  will  permit;  also  the  coupling  transformer  leakage  reactance  and 
resistance  must  not  appreciably  affect  the  impedance  in  series  with  the  grids. 

The  input  requirements  may  best  be  understood  by  referring  to  the  circuit  for  a  class  B 
audio  amplifier  shown  in  Fig.  25,  with  the  various  equivalent  circuit  constants  inserted. 


4-275  +400 

FIG.  25.     Circuit  and  Equivalents  of  a  Class  B  Audio  Amplifier 

The  combination  of  tubes  shown  is  perhaps  the  most  practical  combination  for  power  out- 
puts of  approximately  25  watts.  The  transformer  T  is  merely  an  interstage  voltage  trans- 
former to  supply  audio  voltage  from  the  source  to  the  grids  of  the  type  45  tubes  as  class  A 
amplifiers.  For  the  grid  and  plate  voltages  as  shown,  the  peak  plate-voltage  swing  is 
approximately  165  volts  each  side  of  the  operating  point  for  load  resistances  of  the  order  of 
10,000  ohms,  which  would  be  a  peak  voltage  of  330  volts  on  the  primary  of  transformer  T. 
Since  according  to  Fig.  24  the  grid  swing  for  the  46  tube,  for  full  output  of  about  25  watts, 
is  approximately  55  volts,  the  ratio  of  the  transformer  TI  is  the  ratio  of  330  to  55  or  6  to  1, 
as  measured  from  plate  to  plate  of  the  driver  tubes  to  each  grid  of  the  46  tubes.  This 
transformer  has  an  impedance  ratio  of  36  to  1.  The  plate  resistance,  rp,  of  the  45  tube 
is  approximately  1800  ohms  per  tube  so  that  the  equivalent  resistance  in  series  with  the 


7-18 


VACUUM-TUBE   CIRCUIT  ELEMENTS 


primary  of  TI  is  2rp,  or  3600  ohms  for  the  above  case.    Therefore,  the  equivalent  resistance 
in  series  with  the  grid  of  each  46  tube  is  (in  addition  to  transformer  losses) 


(Turn  ratio)1 


=  n  =  100  ohms 


(18) 


The  actual  impedance  is  usually  about  10  per  cent  higher  than  the  above  calculated 
value  because  of  transformer  losses,  but  such  losses  are  usually  allowed  for  by  assuming  a 
grid  swing  about  10  per  cent  or  more  higher  than  the  actual  swing  needed  for  a  given 
power  output.  In  addition  to  the  equivalent  resistance,  TI,  in  series  with  the  grids  of  the 
46  tubes,  there  is  an  equivalent  inductance,  LI,  which  is  the  equivalent  leakage  reactance 
in  series  with  one  side  of  the  secondary  of  transformer  TI.  As  explained  above,  the  change 
in  input  resistance  of  the  46  tube  is  very  rapid  over  certain  parts  of  the  audio  cycle  which 
results  in  current  flow  to  the  grids  at  harmonic  frequencies  much  higher  than  the  funda- 
mental frequency.  The  frequency  at  which  these  grid  currents  flow  may  be  5  to  10  times 
the  frequency  of  the  fundamental,  depending  upon  the  frequency  and  amplitude  of  the 
signal.  With  currents  flowing  at  such  frequencies  to  the  grids  of  the  46's,  it  is  evident 
that  the  equivalent  inductance  in  series  with  the  grids  due  to  leakage  reactance  of  the 
transformer  TI  must  be  low  in  order  that  the  grid  voltage  at  these  frequencies  will  be 
low.  A  high  leakage  reactance  in  the  input  or  driver  transformer  manifests  itself  in  the 
form  of  a  fuzzy  or  a  ragged  output  wave  over  certain  portions  of  the  cycle,  and  it  may 
cause  very  high  transient  voltages  in  the  plate  circuit,  which  are  responsible  for  many 
output  transformer  breakdowns  and  the  tube  failures. 

OUTPUT  CIRCUIT  REQUIREMENTS.  Assuming  a  low-impedance  circuit  to  the 
grids  of  the  46's,  which  results  in  very  little  distortion  of  voltage  supplied  to  the  grids, 

there  is  still  a  source  of  distortion  in  the 
plate  circuit  because  of  the  deviation  of  the 
plate-current  curves  from  a  straight  line. 
This  deviation  is  shown  in  Fig.  24,  and  if  the 
points  about  the  zero  axis  are  plotted  on  a 
much  larger  scale  and  the  slopes  of  the  two 
plate  currents  added  over  the  period  during 
which  plate  current  flows  to  each  tube,  it 
will  be  found  that  the  per  cent  deviation 
is  perhaps  greater  at  zero  than  at  any  other 
point.  In  the  838  tube  which  was  specially 
designed  for  class  B  operation  there  is  appre- 
ciable plate  current  at  no-signal  condition; 

FIG.  26.  Distortion  Curves  for  RCA  46  Tubes  as  however>  *}*  ^  0*  th*  Plate"current 
Class  B  Audio  Amplifiers  curve  1S  such  that  when  tw°  tubes  are  oper- 

ated in  pushpull  the  resultant  output  current 

is  practically  linear.  This  characteristic  of  the  46  may  be  seen  by  referring  to  the  results  of 
distortion  measurements  (Fig.  26)  as  made  on  an  experimental  amplifier  with  constants 
approximately  as  shown  in  Fig.  25.  The  non-linearity  near  the  zero  signal  is  distinctly 
shown  by  the  rapid  rise  of  the  third  harmonic  at  about  1-watt  output,  after  which  one 
tube  ceases  to  work  for  most  of  a  half-cycle,  resulting  in  a  partial  balancing  out  of  the 
third  harmonic  for  certain  power  outputs. 

From  the  definition  of  the  class  B  amplifier,  and  from  eq.  (17),  it  is  evident  that  the 
plate  current  to  the  class  B  audio  amplifier  tubes  is  approximately  proportional  to  the 
grid  excitation  and  that  the  efficiency  increases  to  a  maximum  at  full  output.  Referring 
to  Fig.  24,  it  is  seen  that,  if  the  plate  voltage  decreases  because  of  poor  regulation  of  the 
plate  supply,  the  peak  plate  currents  will  not  be  reached  as  indicated  by  the  curves. 
Therefore,  the  plate  supply  voltage  should  have  good  regulation  and  should  be  by-passed 
well  for  instantaneous  peak  plate  currents.  (See  p.  7-108.) 

The  shape  of  the  plate  current  curves  in  Fig.  24  cannot  be  used  to  calculate  distortion 
by  the  simple  formulas  given  above  for  the  second  and  third  harmonics.  The  fact  that 
the  output  resistance  from  the  driver  stage  is  about  20  per  cent  of  the  grid  resistance  at 
near  the  peak  swing  alters^  the  plate-current  curve  appreciably  at  the  upper  limits  of 
plate  current,  which  complicates  the  mathematical  determination  of  harmonic  outputs. 
Because  of  the  complicated  nature  of  the  procedure  in  calculating  harmonics,  it  is 'much 
easier  and  more  accurate  (for  small  amplifiers)  to  analyze  the  output  wave,  from  an  ex- 
perimental amplifier  including  the  driver,  for  harmonics  by  means  of  a  reliable  voltage 
analyzer. 

LIGHT-WEIGHT  CLASS  B  AUDIO  AMPLIFIERS.  Because  of  the  high  efficiency 
of  the  class  B  audio  amplifier  its  use  is  very  desirable  in  a  battery  receiver,  in  an  automo- 
bile receiver,  or  where  space  and  weight"  are  limited.  The  use  of  small  tubes  for  a  given 


8         12 
Power  Output  -  ^ 


24 


CLASS  B  AMPLIFIEES 


7-19 


power  output  is  represented  by  a  miniature  6AU6  driving  a  single  1635  class  B  output 
tube.  The  output  power  from  this  combination  is  of  the  order  of  12  to  15  watts  audio 
power  with  low  distortion  and  with  very  low  plate  drain  by  both  the  driver  and  output 
tubes. 

As  explained  above,  bias  for  a  class  B  amplifier  is  difficult  to  obtain  and  keep  constant 
because  of  the  summation  of  grid-current  and  plate-current  peaks  that  must  flow  through 
a  self-bias  resistor  and  because  of  the  grid-current  peaks  if  a  separate  bias  is  used.  There- 
fore the  1635  tube  was  designed  for  zero  bias  at  a  maximum  of  400  volts  plate  supply. 

A  4000-ohm  load  line  may  be  drawn  through  a  point  at  zero  bias  and  400  volts  supply 
to  the  plate  on  a  family  of  published  plate  characteristics  for  the  1635.  The  dynamic 
transfer  characteristics  are  shown  in  Fig.  27  for  a  4000-ohm  load  resistance.  The  grid 


+90 


-90 

35  -30  -25-20   -15   -10    -5        0      +5    +10  +15  +20  -4-25  +30    +35 
Grid  volts 

FIG.  27.     Dynamic  Transfer  Curves  for  a  1635  Tube  as  a  Class  B  Audio  Amplifier 

current  vs.  grid  voltage  is  also  shown  in  Fig.  27.  A  careful  examination  of  the  plate-current 
curve  in  Fig.  27  will  show  that  the  current  is  quite  low  at  zero  bias  but  trails  out  before 
cutoff  similar  to  a  variable-mu  tube. 

At  zero  bias  about  6  ma  plate  current  flows  to  each  plate  so  that  each  plate  contributes 
to  the  output.  Since  the  plate  resistance  of  the  1635  is  high  the  output  power  from  each 
plate  is  measured  by  plate-current  deviation  from  its  zero  value  through  the  load  resist- 
ance. In  pushpull  operation  if  the  plate  current  of  one  tube  is  increasing  the  other  is 
decreasing,  and  since  the  phase  of  the  currents  is  reversed  in  the  output  transformer  the 
output  current  is  given  by  the  difference  in  plate  current  to  the  two  plates.  The  true 
output-current  curve  is  then  obtained  and  plotted  in  Fig.  27  as  the  resultant  output  cur- 
rent. This  resultant-output-current  curve  is  more  nearly  linear  for  the  1635  tube  than 
for  other  double  tubes  because  of  the  special  grid  construction.  Because  of  the  special 
grids  the  grid  current  at  zero  bias  is  lower  so  that  the  sum  of  the  grid  resistance  of  the  two 
tubes  is  higher  than  for  other  double  plate  tubes  during  the  time  grid  current  is  flowing 
to  both  grids.  This  higher  input  resistance  to  the  tube  for  small  signal  swings  reduces  so- 
called  high  frequency  or  "fuzz"  type  of  distortion. 


7-20 


CIECCJIT  ELEMENTS 


At  «  rrawentathr*  P^k  plate  current  of  80  ma  in  Fig.  27  the  output  is  7*Bi»/2  !  -  12.8 
w.t*/  ThT^nd  «ak^g  is  +32  volts  for  the  above  peak  plate  current,  and  the  grid 
™Js  ^a  52.    These  peak  grid  values  of  current  and  voltage  ^permit  the  driver 
i*      The  slope  of  the  grid-current  vs.  grid-voltage  cune  at  +30  tc  '  +3-  volts 
a  •  Stance  of  500  ohms,  increases  to  about  1500  ohms  at  about  +6  volts  on 
d  and  dm^M  again  to  about  700  to  SOO  ohms  near  the  zero  axis  because  of  grid 
*irids.    The  above  variation  of  input  resistance  necessitates  a 


of  the  OAU6  pentode  to  be  used  as  a  driver  is  Quite  high  so  that 
is  used  to  Iow0r  its  output  resistance.    The  circuit  for  the  6ATJ6  driver  and 

the  6AU6  under  the  voltage  ^onditions 

shown  in  Fig  28  it  will  be  seen  that  the  bias  for  class  A  operation  is  about  1.5  volts  at  a 
plat*  mrrent  of  about  7  ma.  At  double  plate  current  and  a  signal  swing  to  zero  bias  the 
plate  *winf>  w  about  225  volts.  A  peak  swing  of  about  32  volts  is  needed  on  the  grids  of 
Uie  1635  BO  that  the  step-down  ratio  of  the  driver  transformer  to  each  grid  is  about  7  to  1. 


6AU6 


1635 


-f-300  -  -{-150  +400 

Ft©,  38»     Drir«r  and  Output  Circuit  for  a  1635  Class  B  Audio  Amplifier 

maximum  mi-rent  output  from  each  side  of  TI  is  then  7  ma  X  7  —  49  ma,  which  is 
than  mm  pie  driving  capability  for  a  needed  peak  grid  current  of  25  ma.  As  noted 
above,  the  variation  of  grid  resistance  or  load  on  T\  was  500  ohms  to  about  1500  ohms. 
Hit  input  roristance  should  be  as  low  as  10  per  cent  of  the  minimum  grid  resistance  to 
keep  the  input  distortion  to  &  low  value. 

Thi?  gm  of  the  6A1T6  is  about  5000  micromhos,  and,  if  it  is  assumed  that  1  volt  peak 
were  applied  to  the  plate  of  the  6ATJ6  and  that  0.2  of  a  volt  peak  gets  back  to  the  grid  of 
til©  6A176  t^mi^a  Jfc,  tfae  effective  output  impedance  of  the  6AU6  to  one  side  of  the 

1  1000 


m       (Ratio  of  m« 

Tfe«  eff«ciir®  imter^J  rc^taaee  of  the  transformer,  TI,  to  each  side  of  its  secondary 
may  be  about  20  ofams  fw  a  good  design  so  that  the  total  input  resistance  in  series  with 
1635  grids  m  alxmit  40  ohms.    This  low  value  of  resistance  will  effectively  reduce  input 
to  a  low  vata*. 

|%.  28  id  the  combined  plate  resistance  of  Vi,  jRs,  and  fij  is  50,000  ohms, 
J£i  in  tJb®  afe®*r«  <^lmla4i(m«  for  feedb&ek  voltage  will  be  200,000  ohms.  Therefore,  20 
par  cemt  01  the  peak  plate  swing  of  225  volts  on  the  plate  of  the  6AU6  will  require  about 
1  ma  of  taw  6AtTt  plate-atrreut  peak  swing  and  apply  45  volts  audio  on  the  plate  of  the 
la  tliis  emse  Ft  EIU^  sup^y  au  equivalent  voltage  of  45  volts  at  its  plate  to 
out  ^e  Iwdlsaek  v^tag®  P*«s  a  peak  voitage  of  about  2  volts  to  drive  the  grid 

k>**i  M  400©  ofeiaa  per  plate  the  plate  to  plate  load  for  Tz  is  16,000 
I  m^sfc  Iiave  tow  leakage  reactance  to  prevent  fuzz-type  dis- 
rtkiFig.25.  A  cliecic  of  constants  for  the  6AU6  as  a  degenerative 
fa*dfca«fc  «var  far  taw  46*1  in  Fig.  25  indieal©s  tfeat  tlie  6AU6  ^  essentially  as  good  a 
4r*fw  m  or  b«t««r  4^®  Ow  two  45T$.  This  indicates  the  effectiveness  of  the  addition  of  a 
•atdfcark  res^w  ^  »  F%.  2S, 

S^ssf  ^«pa«3ri^ioti  nmy  b«  f«d  b«^  from  a  resistance  divider,  R^  across  the  output  of 
Fj  t*»  tht  <«taodt  «i  ^  a«  ^)wa  in  F%.  2S  to  ^crease  tfe©  distortion  of  the  system  further 
Th»  totort»a  vt,  ©utfmi  pow^-  of  tbe  «ira«ifc  dk^ro  ia  Fig.  2S  is  shown  in  Fig  29  The 


CLASS  B  AMPLIFIERS 


7-21 


feedback  for  the  overall  distortion  curve  was  an  amount  required  to  reduce  the  overall 
gain  about  2  or  3  to  1. 

Other  sharp  cutoff  pentodes  such  as  the  6SJ7  may  be  used  as  drivers.  Battery  tubes  of 
the  1.4  volt  series  may  be  used  in  combinations  to  obtain  high  outputs  for  given  input 
powers. 


s  400  volts 
4000  ohms 


6  8  10 

Power  Output,  Watts 


14 


16 


FIG.  29.     Overall  Distortion  of  the  Circuit  of  Fig.  28  Using  the  1635  as  the  Output  Tube 

MEDIUM-POWER  AUDIO  AMPLIFIERS.  A  set  of  curves  similar  to  the  curves  in 
Fig.  24  is  shown  for  the  UV849  tube  in  Fig.  30.  It  will  be  noted  from  these  curves  that 
the  class  B  operating  bias  is  approximately  — 140  volts  and  that  the  instantaneous  slope 
of  the  grid-current  curves  is  positive  and  negative  with  magnitudes  as  low  as  500  ohms. 
Considering  the  relatively  high  grid  voltages  at  which  these  low  instantaneous  grid  re- 
sistances occur,  these  tubes  are  hard  to  drive  without  appreciable  distortion.  However, 
two  UV845-type  tubes  operating  at  full  rated  voltages  may  be  used  to  drive  these  tubes 
successfully  with  an  equivalent  resistance  of  approximately  100  ohms  in  series  with  the 
grids  of  the  UV849  tubes. 


1400 


+40         +80        +120 
FIG.  SO.     Dynamic  Transfer  Characteristics  of  849  Tube 


-160      -120        -80         -40  0 

Grid  Volts 


By  applying  eq.  (17)  to  calculate  the  various  powers  from  data  obtained  from  Tig.  30, 
it  is  found  that  the  maximum  power  output  for  the  2500-ohm  load  line  is  about  1350 
watts  for  two  tubes,  the  plate  loss  is  about  350  watts  per  tube,  with  a  plate-circuit  efficiency 
of  approximately  55  per  cent.  These  calculations  do  not  take  into  account  the  transformer 
losses.  Such  audio  power  outputs  are  used  principally  to  plate-modulate  a  class  C  radio- 
frequency  output  amplifier. 


HIGH-POWER  AUDIO  AMPLIFIER.  The  500-kw  broadcasting  transmitter  operated 
at  WLW  in  Cincinnati,  Ohio,  used  high  level  modulation  with  a  class  B  audio  amplifier 
capable  of  approximately  325-kw  audio  output  power.  At  the  time  this  station  was  first 
operated,  early  in  1934,  it  was  the  world's  largest  broadcasting  station.  The  class  B 
amplifier  for  the  above  station  used  eight  type-862,  100-kw  tubes,  with  12,000  volts  on  the 


As  stated  above,  tubes  other  than  the  ones  discussed  may  be  used  as  class  B  audio 
amplifiers,  but  their  particular  grid  characteristics  must  be  known  in  order  that  a  driver 
system  may  be  properly  designed.  The  load  resistance  and  plate-current  characteristics 
are  also  important  items  to  consider  in  the  design  of  a  satisfactory  class  B  audio  amplifier. 

RADIO -FREQUENCY  AMPLIFIERS.  The  class  B  radio  amplifier  functions  much 
the  same  as  the  class  B  audio  amplifier  as  far  as  the  grid  currents  and  plate  currents  are 
concerned,  except  that  the  plate-current  distortion  may  be  greater  with  a  correspondingly 
higher  output  power  without  serious  signal  distortion.  This  is  because  the  second  and 
higher  harmonics  are  multiples  of  the  radio  frequency  and  are  outside  of  the  frequency 
range  of  interest.  The  principal  difference  between  the  two  amplifiers  is  in  the  input  or 
driver  circuits  and  the  output  circuits.  If  the  input  or  driver  circuit  is  tuned,  the  har- 
monics are  by-passed  and  do  not  appear  on  the  grids,  which  results  in  a  well-regulated  or 
low-impedance  driver  source,  as  far  as  the  fundamental  frequency  is  concerned.  As  in  the 
class  B  audio  amplifier,  the  regulation  of  the  voltage  to  the  grids  of  the  class  B  tubes  de- 
pends upon  the  ability  of  the  driver  tube  to  supply  the  instantaneous  grid  currents  directly. 

Another  factor  which  reduces  distortion  to  a  negligible  value  is  the  fact  that  the  plate 
circuit  is  timed;  thus  it  will  absorb  the  fundamental  component  of  the  plate  current  and 
by-pass  the  harmonics.  The  power  output  of  the  class  B  radio  amplifier  may  be  cal- 
culated like  the  power  output  for  the  class  B  audio  amplifier  if  somewhat  greater  peak 
currents  are  assumed  for  peak  outputs.  It  would  seem  from  the  above  that  the  class  B 
radio  amplifier,  also  known  as  the  linear  amplifier,  is  inherently  a  simple  type  of  amplifier 
to  design;  this  is  true  if  it  is  desired  only  to  obtain  a  power  amplification  of  the  funda- 
mental frequency.  However,  it  is  not  economical  to  use  such  an  amplifier  as  merely  a 
tuned  amplifier  for  voltage  or  power  amplification,  because  in  such  cases  the  class  A 
amplifier  would  probably  be  better  for  voltage  amplification  and  the  class  C  amplifier  for 
power  amplification.  The  real  use  for  the  class  B  radio  amplifier  is  in  circuits  where  a 
low-powered  modulated  signal  is  to  be  amplified  to  a  higher  power  level.  Referring  to 
the  definition  of  the  class  B  amplifier,  one  characteristic  of  the  amplifier  is  that  the  output 
voltage  is  proportional  to  the  input  voltage;  therefore,  if  a  modulated  radio-frequency 
signal  is  used  to  excite  the  grids  of  a  class  B  radio  amplifier,  the  output  for  an  ideal  case 
would  be  a  modulated  signal  identical  to  the  modulated  input  signal,  except  that  it  would 
be  at  a  much  higher  power  level.  The  increase  in  power  level  is  20  to  100  times,  depending 
upon  the  degree  to  which  the  amplifier  is  driven. 

A  typical  circuit  for  such  an  amplifier  is  shown  in  Fig.  31,  in  which  the  tuned  input 
circuit  LiCiCzCs  is  so  arranged  that  C%  equals  Cz  and  each  has  a  value  with  respect  to 


Bta-s  far  Approximate^*1 
Plate  Current  Cufcotf 

FIG.  31.    Circuit  for  Class  B  Radio  Amplifier 

Ci  that  permits  the  desired  grid  swing  of  the  tubes,  Vi  and  72,  for  a  given  modulated  sig- 
nal applied  to  the  input  circuit.  Inductances  L2  and  L3  are  radio-frequency  chokes  to 
provide  a  low  d-c  resistance  path  for  the  grids,  and  rx  and  rz  are  resistors  from  the  grids 
to  ground  to  suppress  undesirable  oscillation.  C6  and  C7  are  neutralizing  condensers, 
and  the  output  tuned  circuit  is  LtdCs,  in  which  C4  and  C5  are  equal.  The  radio-frequency 
choke  1/5  provides  a  floating  center  tap  for  L4  because  the  tuned  circuit  is  center-tapped 
in  the  condenser  branch.  The  circuit  as  shown  is  for  a  pushpull  class  B  radio-frequency 
amplifier,  but,  as  stated  above,  a  single  tube  may  be  used  in  a  circuit  similar  to  the  one 
discussed,  by  leaving  out  the  unnecessary  parts  or  sections  of  the  circuit.  The  type  of 
circuit  shown,  or  its  equivalent,  was  in  common  use  in  most  broadcasting  stations  of  1000 


CLASS  B  AMPLIFIEKS 


7-23 


watts  or  more.    However,  the  present  trend  is  toward  the  use  of  the  more  efficient  high 
level  modulation  system.    (See  p.  7-74.) 

Distortion.  As  discussed  above,  there  is  little  distortion  from  a  tuned  input  and  tuned 
output  amplifier  as  far  as  the  fundamental  radio  frequency  is  concerned.  Since  the  prim- 
ary use  of  the  linear  amplifier  is  to  amplify  modulated  signals  without  distortion,  the 
linearity  of  the  amplifier  becomes  a  factor  which  determines  almost  entirely  whether  the 
demodulated  output  is  distorted  or  not.  The  linearity  of  such  an  amplifier  is  determined 
by  the  ratio  of  output  current,  or  voltage,  against  rms  input  voltage.  The  curves  of 
Fig.  32  are  plotted  from  actual  experimental  data  obtained  on  two  RCA  846  tubes  in 
pushpull  as  a  class  B  radio  or  linear  amplifier  with  7500  volts  on  plate.  For  a  bias  of 
—  150  volts,  the  output  tank  current  is  approximately  linear  with  respect  to  the  grid- 
excitation  voltage  from  zero  to  about  4  amp.  If  the  amplifier  is  used  as  a  radio-frequency 
output  amplifier  in  which  the  input  signal  is  100  per  cent  modulated,  the  normal  output 


' 


E&  g=s  7BOO  Vblts 
-  r=6000  Ohms  per  Tube          - 
Output  Resistance  =» 374  Ohms 


400  6OO  800 

R,P.  etfd  Volts  perTobe  CR.M.S.) 

FIG.  32.     Characteristics  of  RCA  846  as  Class  B  R-f  Amplifier 

tank  current  would  be  approximately  2  amp,  which  corresponds  to  a  grid  excitation  of 
470  volts  for  approximately  1300  watts  output.  It  will  be  noted  that  such  high  per  cent 
modulation  of  the  excitation  voltage  in  this  case  causes  a  deviation  from  a  straight  line 
at  both  the  upper  and  lower  limits  of  swings  so  that  the  predominating  harmonic  in  the 
demodulated  output  signal  is  the  third.  If  a  lower  bias  is  used  the  curve  would  be  more 
nearly  linear  from  the  operating  point  of  2  amp,  but  the  slope  at  the  upper  end  would  be 
approximately  the  same  resulting  in  a  second  harmonic  for  100  per  cent  modulation. 

The  flattening  of  the  output  tank  current  near  the  zero  axis,  for  the  curves  in  Fig.  32, 
is  due  to  the  large  bias  chosen;  however,  the  flattening  of  the  upper  end  of  the  curve 
may  be  due  to  emission  or  space-charge  limitations  which  limit  the  peak  plate  currents. 
Another  factor  that  often  contributes  to  the  flattening  of  the  upper  end  of  the  output 
tank  current  curve  is  the  inability  of  the  modulated  exciter  to  supply  the  higher  value 
of  average  grid  current  without  distortion  to  the  output  tubes  at  the  peak  values  of  grid 
excitation.  The  driver  must  have  low  impedance  to  the  grids  of  the  output  tubes  to 
prevent  distortion,  for  practically  the  same  reasons  as  a  low  impedance  is  required  for  the 
input  to  the  class  B  audio  amplifiers. 

Since  the  normal  carrier  value  of  output  current  for  the  class  B  radio  amplifier  is 
approximately  one-half  the  peak  value  and  occurs  at  one-half  the  peak  value  of  grid  excita- 
tion, it  can  be  seen  from  eq.  (17)  that  the  efficiency  of  the  plate  circuit  is  relatively  low. 
(See  p.  7-17.)  This  efficiency  is  usually  20  to  30  per  cent.  The  carrier  power  output  is 
only  25  per  cent  of  the  peak  power  output;  therefore,  except  in  tubes  where  peak  plate 
voltage  or  peak  plate  power  input  is  limited,  four  times  as  many  tubes  are  theoretically 
required  for  class  B  radio  amplifiers  for  a  given  power  output  as  are  required  for  a  class  C 
high  level  modulated  radio-frequency  amplifier.  Actually  the  water-cooled  tubes  are 
limited  by  peak  plate  voltage  for  the  plate  modulated  class  C  amplifier  so  that  the  number 
of  tubes  for  a  class  C  amplifier,  when  these  tubes  are  used,  is  approximately  one-half  that 
required  for  a  class  B  radio  amplifier. 


7-24  VACUUM-TUBE  CIRCUIT  ELEMENTS 

SUMMARY  IOB.  CLASS  B  AMPLIFIERS.  Some  of  the  more  important  require- 
ments to  meet  in  the  design  and  operation  of  class  B  amplifiers  and  some  of  their  charac- 
ter istk$  may  be  summarized  as  follows. 

1.  The  rapiit  system  to  the  class  B  audio  amplifier  must  have  low  impedance,  especially 
iC  the  instantaneous  values  of  grid  current  change  rapidly  either  negatively  or  positively. 
The  dbp©  of  the  grid-current  curve  or  its  rate  of  change  per  volt  applied  to  the  grid  is  the 
grid  mastaoe  at  the  point  taken.    The  equivalent  inductive  reactance  in  series  with 
each  grid  because  of  driver  transformer  leakage  is  a  part  of  the  impedance  in  series  with 
the  grids;  it  must  be  kept  very  low  if  the  instantaneous  values  of  grid  resistance  are  low 
aad  change  rapidly  over  the  audio  cycle. 

2.  Failure  to  meet  the  low-impedance  input  requirements  usually  results  in  appreciable 
distortion,  and  generally  the  output  transformer  is  subjected  to  high  peak  voltages  which 
may  damage  the  transformer  or  the  tubes. 

3.  The  load  resistance  the  tube  works  into  is  one-fourth  of  the  calculated  plate  to  plate 
impedance  of  the  loaded  output  transformer.    This  applies  to  radio-frequency  as  well  as 
audio-frequency  amplifiers. 

4.  The  load  line  for  the  class  B  amplifier  for  radio  and  audio  frequencies  should  be  as 
high  a  resistance  as  possible  consistent  with  grid  current  peaks  and  space-charge  limita- 
tion in  order  that  the  plate  efficiency  may  be  maximum.    Somewhat  higher  peak  plate 
currents  may  be  assumed  for  the  class  B  radio  amplifier,  because  of  the  tuned  plate 
circuit, 

5.  The  class  B  audio  amplifier  is  the  most  efficient  type  of  amplifier  for  audio  fre- 
quencies, and  may  attain  a  plate  efficiency  of  65  to  70  per  cent  at  full  output  power  for 
some  of  the  larger  tubes. 

0.  Hi®  dase  B  radio  amplifier  has  poor  plate  efficiency,  which  is  usually  20  to  30  per 
eeat,  so  ihsA  a  rel&ti'roly  liigh  tube  power  capacity  is  required  for  a  given  output. 

7,  Tbe  ciaas  B  radio  amplifier  also  requires  a  well-regulated  or  low-impedance  driving 
scmr«je,  but  the  re<|uir«Daents  are  not  as  strict  as  the  driver  requirements  of  the  class  B 
audio  amplifier,  because  the  radio  amplifier  may  have  a  tuned  input  circuit, 

S»  The  plate  supply  for  the  class  B  audio  amplifier  must  have  good  regulation,  espe- 
cially if  tubes  witfe  higa  bias  are  used.  The  plate  supply  regulation  is  not  so  important 
with  the  class  B  radio  amplifier  because  the  average  plate  current  is  essentially  constant. 
Btifficieiit  plate  supply  by-pass  capacitance  is  necessary  in  either  amplifier  to  maintain 
constant  plate  voltage  over  the  audio  cycle. 

3.  CIASS  C  AMPLIFIERS 

GENERAL.  Th®  class  C  amplifier  by  definition  is  an  amplifier  that  is  biased  con- 
s«teraMy  beyond  plate  current  cutoff,  so  that  plate  current  flows  for  a  period  less  than 
!$@  electrical  degrees*  during  the  time  the  grid  swings  in  a  positive  direction  from  its 
twraud  v»ls*&  of  bias  potential.  In  general,  the  grid  is  driven  to  the  point  of  saturation, 
thftt  15,  to  a.  point  m&  wfeieti  the  output  voltage  is  no  longer  proportional  to  input  voltage. 
Uncles-  tlo»  0®ndffcitM  the  efficiency  of  tlie  amplifier  is  approximately  maximum  for  any 
gjifTMa  piAtft^-ciresiBt  conditions,  the  output  voltage  is  proportional  to  the  d-c  plate  voltage, 
and  ike  oatpat  power  is  proportional  to  the  square  of  the  d-c  plate  voltage. 

Bm&e*  in  gMMral,  appreciable  power  is  required  to  drive  the  class  C  amplifier  because 
of  tli®  rdtowiy  tai«Ji  positive  grid  swings,  this  type  of  amplifier  is  essentially  a  power- 
atsiptifier  dwnm.  Because  of  the  distorted  plate  current  that  flows  during  only  a  part 
of  &  Imlf-cyelfe  H  m  uteessary  to  tuae  the  output  or  plate  circuit  of  the  amplifier  if-  an 
B®«te0rt*d  output  WVF®  is  tiesred.  Therefore,  the  class  C  amplifier  is  essentially  a 
«t^»-irftqiiw<^  amplifier  aad  is  used  as  the  output  device  for  radio  telegraphy  or  high 
3*w4  Eaoduiat*d  tetephony  transmitters. 

CIBCIHT  CALCULATIONS,    Since  the  class  C  amplifier  can  be  used  only  as  a  tuned 
pi  wfe*r$  distortion  of  the  output  wave  is  permitted,  it  is  obvious  that 
Ms  ampliSer  M  nmefe  more  limited  taan  the  applications  of  the  class  A 
m  wsapiiiwrs.    Im  Hie  dbas  C  amplifier  it  is  diScuit  to  predict  the  performance  of 
ftwail»^»t^«i«racl«rWcs.    As  long  as  the  plate  circuit  load,  r,  has  a  sufficiently 
t©  limit  the  miaimiim  ia^miaaeoiK  plate  voltage  to  a  value  such  that  the 
9M*»  aimo*  m  «**nti«J}y  ^|f  sine  waves,  ike  power  output  and  other  characteristics  of 
WftOTOHt  wd  tote  may  be  eafcmlaied  la  mudi  the  same  manner  as  for  the  class  B  audio 
Hw»w,  umW  these  conditions,  the  piate-drcuit  efficiency  is  usually  less 
«mt.    Wtai  th*  load  msstaace  is  high  tbe  minimum  instantaneous  plate 
****^  *rs*WW3sfii  •**  »  th«*  essentially  no  plate  current  can  flow  at  the 
,  **K-*U-*>  th*  pfcto  Y^t^  »  ^s»r  aero. 


CLASS  C  AMPLIFIERS 


7-25 


To  illustrate  better  the  above  features  of  the  class  C  amplifiers  as  well  as  the  general 
circuit  features,  it  is  well  to  discuss  the  typical  class  C  amplifier  circuit  such  as  shown  in 
Fig.  33.  The  proper  constants  to  use  in  the  circuit  depend  upon  the  tube  used,  general 
class  of  service,  and  the  values  of  maximum  grid  and  plate  currents  allowed.  For  the 
204A  tube  the  maximum  allowed  constants  are:  d-c  plate  voltage,  2000  volts;  d-c  plate 
current,  275  ma;  plate  dissipation,  167  watts;  d-c  grid  current,  SO  ma. 

If  a  modulation  factor  of  1  is  applied  to  the  plate  voltage,  it  is  evident  that  at  peak 
upward  modulation  the  instantaneous  plate  voltage  will  be  4000  volts.  In  order  that 
plate  current  cutoff  may  be  obtained  at  4000  volts  on  the  plate,  so  that  the  tube  may 
continue  to  operate  as  a  class  C  amplifier  at  this  voltage,  the  bias  required  is  approximately 
—  200  volts,  but  a  bias  somewhat  beyond  plate-current  cutoff,  such  as  —250  volts,  should 
be  used.  Therefore,  since  the  allowable  direct  current  through  the  bias  resistor,  rjT  of 
Fig.  33,  is  a  maximum  of 
SO  ma,  a  voltage  drop  of  «JL 

250  volts  requires  a  resist-     • — L- 

ance  of  3125  ohms  for  TI. 
The  power  dissipated  in 
n  is  approximately  20 
watts,  and  the  power  dis- 
sipated on  the  grid  of 
the  tube  is  comparatively 
small  because  of  the  low 
resistance  of  the  grid  when 
the  grid  current  is  large. 


FIG.  33.     Typical  Circuit  for  Class  C  Amplifier 


If  it  is  assumed  that  one-third  of  the  bias  power  is  dissipated  on  the  grid,  the  total  power 
required  of  the  r-f  input  circuit  is  approximately  30  watts. 

This  power  must  be  supplied  by  the  exciter  amplifier,  and  the  loss  in  the  resistor  TI 
cannot  be  supplied  by  means  of  a  separate  voltage  supply  in  the  hope  that  the  exciter 
is  required  to  supply  the  grid  losses  onlj-.  The  grid-leak  method  of  supplying  bias  as 
shown  is  a  self-biasing  arrangement  and  consumes  no  power  from  the  plate  supply.  The 
one  disadvantage  of  this  method  of  supplying  grid  voltage  is  that  if  the  excitation  is  lost 
the  class  C  tubes  have  no  bias,  so  that,  in  the  case  of  the  204A,  the  resulting  plate  current- 
would  cause  excessive  plate  dissipation.  This  difficulty  may  be  removed  by  using  a  com- 
bination of  a  fixed  bias  (or  biasing  resistor  in  the  cathode  to  negative  plate  supply  lead) 
to  limit  the  plate  current  if  the  excitation  is  lost  and  the  additional  bias  supplied  by  a 
leak  system  as  shown. 

The  input  circuit  to  the  tube  is  tuned  and  the  inductance,  LI,  is  center-tapped  in  order 
that  the  amplifier  may  be  neutralized  by  means  of  the  condenser  Cs.  The  plate  circuit 
is  tuned  by  means  of  €4  and  La,  which  in  turn  is  coupled  to  an  output  circuit.  The  output 
circuit  reflects,  to  the  C±Lz  circuit,  an  equivalent  resistance  of  2r  across  the  tuned  circuit. 
Therefore,  r  is  the  approximate  resistance  the  tube  works  into  during  the  time  plate  cur- 
rent flows. 

POWER  CALCULATIONS.  The  efficiency  of  the  plate  circuit  of  the  class  C  amplifier 
can  be  estimated  quite  accurately  if  the  peak  a-c  voltage  across  the  tank  circuit  C^L* 
is  measured  and  used  in  eq.  (17),  as  for  the  class  B  amplifier,  provided  the  plate  current  is 
approximately  one-half  sine  wave  or  that  the  fundamental  component  of  the  current  is 
very  large  compared  to  the  harmonic  components.  For  this  case  it  will  be  seen  that  the 
expression  /max  ?"  is  in  reality  the  peak  a-c  voltage  measured  across  the  plate  tank  circuit 
in  which  Jmax  is  the  peak  value  of  the  fundamental  component  of  plate  current  and  r 
is  the  value  of  resistance  the  tube  works  into  during  the  time  plate  current  flows.  The 
values  of  Jmax  and  r  are  more  or  less  fictitious,  but  their  effective  product  is  approximately 
the  measured  peak  plate  voltage. 

Referring  to  the  ratings  on  the  204A  as  given  above,  the  permissible  input  plate  power 
is  550  watts  with  only  167  watts  plate  dissipation,  which  requires  a  plate  efficiency  of 
approximately  70  per  cent.  Therefore,  from  eq.  (17),  it  is  found  that  the  peak  voltage 
across  the  tank  circuit  is  1778  volts.  This  peak  voltage  leaves  only  222  volts  on  the  plate 
to  cause  the  peak  plate  current  to  flow.  The  power  output  for  the  above  power  infmt  and 
plate  loss  is  383  watts,  and  if  the  peak  plate  voltage  swing  is  177S  volts,  as  calculated 
above,  the  approximate  value  of  2r  is  found  to  be  4150  ohms,  which  is  tbe  eqtuvaieiit 
load  resistance  across  the  plate  tank  circuit,  and  the  resistance  the  tube  works  into  is  r, 
or  approximately  2075  dims.  The  peak  value  of  the  fundamental  component  of  plate 
current,  as  found  from  the  expression  for  peak  a-c  plate  voltage,  is  860  ma.  It  is  doubtful 
if  this  value  of  plate  current  Sows  at  the  instant  the  plate  voltage  readies  a  roinmmm 
vahie  of  222  volts.  However,  plate-cimiit  efficiencies  of  70  per  cent  are  easily  attained, 
so  that,  even  though  the  actual  plate  current  may  be  considerably  below  S60  ma  at  the 


7-26 


VACUUM-TUBE  CIKCUIT  ELEMENTS 


instant  the  plate  voltage  is  222  volts,  the  peak  value  of  the  fundamental  component  may 
be  SaO  ma  without  sw^dent  plate-current  distortion  to  affect  the  plate  loss  seriously. 

Exmmtic*?  has  shown,  however,  in  experimental  class  C  amplifiers  that  approximately 
70  to  75  per  cent  piat«s-tirciiit  efficiency,  for  a  circuit  as  shown  in  Fig.  33,  is  the  limit  of 
efficiency,  because  the  actual  plat®  current  for  higher  load  resistances  deviates  so  far  from 
a  half  sine  wave  of  current  at  higher  peak  values  of  a-c  plate  voltage  that  the  plate-current 
tjanaonic!  Icwae*  increase  to  keep  the  plate  efficiency  at  a  more  or  less  constant  value,  in 
«©!**  &f  the  f*<*  that  the  measured  peak  a-c  plate  voltage  increases. 

EPSCUU,  FILTER  FOE  HIGH  EFFICIEHCY.  Equation  (17)  indicates  that  if  an 
«3aeati*l!y  feaH  we  wave  of  plate  current  can  be  caused  to  flow  to  the  plate  the  efficiency 
mmy  *»  <^n»eci  to  increase  as  the  peak  tank  \-oltage  increases. 

When  tfe*  P«»k  *»«*  voltage  approaches  the  plate  voltage  the  efficiency  by  eq.  (17) 
emrmxfam  78.5  per  cent.  If  the  peak  plate  voltage  could  reach  1,27  times  the  d-c  plate 
voltage,  the  efficiency  would  be  100  per  cent,  provided  that  the  plate  current  is  essentially 
a  half  ^m  wave  and  the  plate  resistance  of  the  tube  during  the  time  current  flows  is^zero. 
This  100  per  cent  efficiency  condition  is  impossible,  of  course,  but  the  limit  of  efficiency 
is  flee*  to  be  100  per  cent,  because  the  plate  circuit  is  tuned  and  filters  may  be  inserted  in 
news  with  the  plat«  to  prevent  appreciable  flow  of  harmonic  plate  currents.  Such  a 
filter  circuit  is  shows  in  Fig,  34.  The  inductance  L*  is  comparatively  large  and  offers  a 

high  impedance  to  the  fundamental  and  har- 
monic plate  currents.  The  primary  function 
of  LI  is  to  provide  a  path  for  d-c  plate  cur- 
rent but  to  offer  a  high  impedance  to  all  a-c 
components  of  the  plate  current.  The  induct- 
ance In  and  capacitance  Cs  are  tuned  to 
series  resonance  for  the  fundamental  fre- 
quency of  the  circuits.  Therefore,  the  band- 
pass filter  offers  low  impedance  to  the  funda- 
mental component  of  plate  current  but  is 
designed  to  offer  high  impedance  to  harmonic 
components  of  the  plate  current. 
The  value  of  tfee  more  or  less  fictitious  plate  resistance  is  a  function  of  the  tube  and  its 
evfHwtkm,  and  it  may  be  calculated  if  ail  the  constants  of  the  tube  and  circuits  are  known. 
Howtwr,  an  apparent  resistance  of  the  tube  may  be  easily  calculated  from  the  plate 
efficiency.  This  is  done  by  assuming  that  the  apparent  load  resistance  is  in  series  with 
an  apparent  plate  resistance  that  will  give  the  measured  or  calculated  plate-circuit  effi- 
cieacy. 

Such  a  formula  is  given  below: 


Circuit   for   Iiaereased 
Data  C  Amplifier 


—  . 


—  2r 


apparent  plate  resistance 


(19) 


in  wMch  2r  is  the  equivalent  resistance  in  parallel  with  the  tank  circuit  due  to  the  load 
aoiaiiertecl  to  the  class  C  araplilier.  Since  the  tube  functions  for  approximately  one-half 
tJbe  tine*  th®  actual  effective  resistance  of  the  tube  is  approximately  one-half  of  the  ap- 
parent neiistaiHse  ?*&,  and  the  load  the  plate  works  into  is  one-half  the  equivalent  load 
ikee  or  r.  The  load  resistance  r  as  calculated  for  the  204A  at  rated  conditions  as 
atxrre  is  ^07S  ohms,  and  the  calculated  efficiency  is  70  per  cent.  Substituting  these 
i  in  formula  <lf)t  the  effective  resistance  of  the  204A  is  about  890  ohms. 
If  the  harm«mie  coropofwjnts  o€  the  plate  current  are  to  be  effectively  reduced,  the  im- 
peskufje  ©f  the  draft  -W*r*,  when  tuned  to  the  fundamental  frequency,  must  be  4  or  5 
twaes  the  approximate  e&eetiv©  resistance  of  the  tube  at  the  second  harmonic  of  the  funda- 
mwrta!  fntqpMmx*  Tbe  impedance  to  higher  harmonics,  of  course,  will  be  still  higher. 
TliwdbTt,  a  ttmgh  approximation  of  what  the  impedance  of  the  filter  should  be  at  the 
•MMMl  imn«*©i8ie  m  «*|mal  to  the  normal  effective  resistance  across  the  output  tank  circuit 
Kfc  M  power  output  of  the  tube. 

S^^i  imp^iinw  at  the  second  tmrasonic  frequency  necessitates  a  comparatively  large 
I*  t»  C  m&®  lor  the  series  toned  circuit  £4C&-  Since  the  a-c  component  of  plate  current 
iwiMft  «lra*fr  this  ieries  &in*d  circuit,  tlie  voltage  across  Li  and  C6  reaches  quite  high 
wiiMB  M  the  fmii4»3s*i*fea]  fre<t*iesey,  and  tis^e  voltage  across  L3  at  all  the  harmonic  frequen- 
cies M  «onp«rtttiTriy  fa^s.  However,  the  system  permits  approximately  only  half  sine 
waeis  erf  pb*«  curmst  at  tfe«  teidameiital  frequency  so  that  high  efficiency  can  be  achieved. 
is»  type  «£  filler  Isa®  be*s  succsessfuily  xtsed  to  increase  materially  the  efficiency  of  a 
«C  Mttplifiiiri iathe  kfocwalc^y,  aaid  was  raceeeBftxDy  ap^Ued  to  at  least  one  broadcasting 
tfe*  c^rc^rt  AM  i*o4  be^a  used  extensively  in  transmitters  to  date.  The 
on  piat*  lo^,  power  output,  and  e^casney  of  a  laboratory  test  on  a 


CLASS  C  AMPLIFIERS 


7-27 


640 


180 


SO 


10OO  2QOO        4QOO 


6000          8000        ,1.0000 
Etaie  Circuit  Load  Bejstslance 


[45 

12JOOO       14000 


FIG.  35.     Performance  of  RCA  204A  as  Class  C  Amplifier 


204A  is  given  in  Fig.  35.  It  will  be  noted  that  for  low  load  resistances,  that  is,  resistances 
that  limit  peak  a-c  plate  voltages  to  values  appreciably  below  the  d-c  supply  voltage,  the 
filter  adds  little  if  any  to  the  efficiency  of  the  plate  circuit.  This  is  due  to  the  fact  that 
heavy  loads  cause  essentially  one-half  sine  waves  of  plate  current,  so  that  there  are  prac- 
tically no  harmonic  components  of  plate  current  to  reduce. 

The  results  of  the  experimental  data  as  plotted  in  Fig.  35  indicate  that  the  efficiency 
of  a  class  C  amplifier  without  a  filter  is  essentially  constant  over  practically  all  the  load 
range  that  might  be  used.  The  204A  tube,  with  which  the  data  for  the  above  curves  were 
taken,  was  operated  at  approximately  optimum  grid  excitation  so  that  the  values  read 
from  the  curves  represent  approximately  the  best  performance  of  the  tube.  Any  loss  in 
the  filter  for  the  curves 
in  Fig.  35  is  included  as 
plate  loss,  so  that  the 
plate  efficiency  is  actually 
greater  than  shown  when 
the  filter  was  used. 

Oscillograms  of  the  ac- 
tual plate  current  in  a  low- 
frequency  class  C  ampli- 
fier  were  made  on  an  ex- 
periment al  amplifier ;  they  o  ^  °  " 
substantiate  the  above  ex- 
planation of  the  behavior 
of  the  filter  and  indicate 
quite  definitely  that  the 
plate  current  to  the  tube 
at  most  practical  load 
resistances  is  a  double 
peaked  affair,  whereas 
with  the  filter  the  plate 
current  is  essentially  one- 
half  of  a  sine  wave.  As  a  result  of  the  peaked  condition  of  the  plate  current  without 
a  filter,  the  peak  emission  required  from  the  filament  of  the  tube  in  a  class  C  amplifier 
is  6  to  12  times  the  d-c  value;  when  a  filter  is  used  the  peak  emission  required  is  one-third 
to  one-fifth  of  the  peak  without  the  filter.  The  reduced  peak  plate  current  requirement 
when  the  filter  is  used  is  a  decided  benefit. 

Although  the  applications  of  the  class  C  amplifier  to  radio  devices  are  limited  in  num- 
ber, it  is  the  most  efficient  type  of  vacuum-tube  amplifier  known.  When  the  class  C 
amplifier  is  used  as  the  output  system  for  a  radiophone  transmitter,  in  which  a  class  B 
audio  amplifier  is  used  to  modulate  the  plate  supply  to  the  class  C  amplifier,  a  radiophone 
transmitting  station  is  obtained  that  has  a  greater  overall  efficiency  than  any  other  type 
of  transmitter  now  in  general  use. 

SUMMARY  FOR  CLASS  C  AMPLIFIERS.  Some  of  the  more  important  circuit  re- 
quirements of  the  class  C  amplifier  as  well  as  some  of  the  more  important  operating 
points  are  summarized  below: 

1.  If  full  output  is  to  be  obtained  from  the  class  C  amplifier,  the  grid  excitation  should 
be  sufficient  to  obtain  essentially  full  permissible  d-c  grid  current  at  the  recommended 
bias. 

2.  The  grid  excitation  for  a  plate-modulated  class  C  amplifier  must  be  sufficient  to 
cause  a  linear  relation  between  the  plate  voltage  and  output  voltage. 

3.  In  a  plate-modulated  class  C  amplifier,  it  must  be  remembered  that  the  peak  plate 
voltage  due  to  modulation  peaks  is  double  the  plate  voltage  for  normal  carrier,  which  in 
turn  doubles  the  average  plate  current  and  also  the  peak  current  corresponding  to  a 
power  four  times  the  carrier  power.    Therefore,  it  is  necessary  to  determine  whether  the 
tubes  used  in  such  an  amplifier  are  capable  of  withstanding  the  peak  d-c  plate  voltage  over 
an  audio  cycle  and  whether  the  emission  of  the  tube  is  ample  to  meet  the  peak-plate-current 
conditions.  .    . 

4.  Failure  to  meet  the  peak-plate-current  condition  in  item  3,  because  of  emission 
limitation  or  insufficient  excitation,  results  in  a  second  harmonic  in  the  detected  carrier 
output.    Failure  to  meet  the  requirements  in  item  3  only  reduces  the  power  output  and 
efficiency  in  the  case  of  class  C  amplifiers  for  telegraphy. 

5.  The  bias  for  a  class  C  amplifier  must  be  beyond  plate-current  cutoff  at  the  highest 
instantaneous  d-c  plate  voltage.    For  100  per  cent  modulation  the  bias  should  be  2.5  to  3 
times  the  bias  for  plate  current  cutoff  at  normal  plate  voltage.    Excessive  bias  usually 
results  in  higher  excitation  power  with  little  gain  in  efficiency. 


7-28  VACUUM-TUBE  CIRCUIT  ELEMENTS 

0.  The  plate  dissipation  of  a  tube  in  a  class  C  amplifier  is  determined  entirely  by  the 
plat©  fupply  voltage  and  the  load  resistance  for  normal  class  C  operation  of  the  tube,  and 
the  value  0!  load  resistance  has  essentially  no  relation  to  the  plate  resistance  of  the  tube. 

7.  Tbe  ma^tiitiwrn  plate  efficiency  of  a  class  C  amplifier  is  approximately  70  to  75  per 
cvnt  for  all  practical  load  resistances.  This  efficiency  may  be  raised  to  SO  or  90  per  cent 
fey  wring  somewhat  higher  resistance  plate  loads  if  a  band-pass  filter  is  used  hi  series  with 
the  plate, 

&.  Hie  hsoid-pass  filter  also  reduces  the  peak-plate-current  requirements,  which  tends 
to  rtdnae  dfafeortkm  at  high  percentage  modulation. 

0.  Tfefc  audi-o  power  required  to  modulate  the  plate  circuit  is  50  per  cent  of  the  amplifier 
input  plat*  power;  therefore,  any  increase  in  plate  efficiency  reduces  the  modulator  power 

&  REGENERATION  AND  ITS  PREVENTION 

COHDITIOH S  FOR  REGENERATION.  In  each  of  the  theoretical  amplification  equa- 
tkms  feq*.  [4],  [6],  etc.)  the  internal  input  impedance  Z&  appears.  When  rt-s  is  a  positive 
quantity  the  greatest  amplification  will  be  obtained  by  making  the  internal  input  impe- 
dance as  great  as  possible.  However,  it  is  possible  for  r,-2  to  have  a  negative  value  (p.  5-49) 
when  the  external  plate  impedance  of  the  succeeding  tube  is  inductive.  In  this  casa  the 
conditions  uirader  which  maximum  amplification  will  be  obtained  are  quite  different. 

If  rs  U  segmtive  and  kss  than  twice  the  magnitude  of  the  resistance  presented  to  the  grid 
filament  terminals  by  the  rest  of  the  input  circuit,  it  is  possible  for  the  amplification  to  be 
fr*a£er  than  the  value  obtained  when  the  Internal  input  impedance  is  infinite,  for  the  plate 
cwnfet  of  a  single  tube  "m  terms  of  the  impressed  voltage  is  (see  eq.  [1],  p.  7-03) 

(20) 


r?y(?t't  jx-^i 

n  4-  r,  4-  ifa  -f  *,)  i.      r,  -I- 1  \ 


(20a) 

If  either  r»  or  £*  is  infinite,  the  first  factor  reduces  to  tie  ft  ,  but  if  r,-  is  negative  and  less  than 
twice  tlie  magnitude  of  re  (and  x»  and  x«  are  of  opposite  sign  and  their  sum  less  than  x»)  the 
first  factor  will  be  greater  than  jur'f.  The  tube  is  then  said  to  be  regenerating.  In  particu- 
lar when  the  denominator  is  sero  the  amplification  will  be  infinite  and  the  tube  will  oscillate; 
thai  is,  plate  current  of  a  particular  frequency  will  flow  even  when  there  is  no  externally 
iS3apra»d  aJtematinu  voltage  on  the  grid. 

In  an  impedasMse-capitatance  coupled  amplifier  (eq.  [6],  p.  7-04)  regeneration  occurs 


Wl»ii  tranricnarix^eoapletl  amplification  is  need  and  the  circuit  is  properly  tuned  (eq. 
P*I>  ra«&i*®rm&ls$i  wsciira  when  0  >  rif  >  2r^  and  oscillation  when  r&  =  ~r4. 

Hie  eetttwrirnt  mg@£ift  reti^aiicr  of  the  input  circuit  is  caused  by  the  plate  voltage 
*foetrc*t*tiaa%  iodoci»K  a  voltage  between  the  grid  and  filament  which  has  a  component 
with  tfe*  imiweaeed  grid  voltage.  This  will  occur  only  when  the  plate  load  is 
.  (B©e  p.  5-49.)  A  similar  voltage  impressed  on  the  grid  by  any  other  means 
of  www,  haw  the  same  effect.  (See  Regeneration,  Theory  and  Experiment,  by 
«m,  Kiwr,  and  Ware,  Free.  LKJ8.,  October,  19S5.)  Circuits  are  frequently  designed 
m  thai  pert  <rf  the  *mrfy  m  the  plate  circuit  is  "fed  back'*  to  the  grid  circuit  to  increase 
the  tduplifieation.  When  the  feedback  is  inductive  through  a  movable  coil,  this  coil  is 


. 

1IW1CTS  OF  UGnVKATIOH.  The  increased  amplification  obtained  as  a  result 
^  i«»f««mtwi  is  gc^^etiims®  w&d  t©  iBcreafe  gain,  partieulaxly  where  a  single  frequency  is 
oetat  a«pitei  Smae  a  r«®isiaerativ0  csiraiit  usiially  behaves  like  a  very  sharply  tuned 
««wit  it  is  not  useful  when  &  broad  band  of  frequencies  must  be  amplified. 

E^w»r»ii«i  tftrtito  in  a  dewaee  is  th®  elective  mfsit  resistance  of  the  amplifier  and 
eo  tfamwi!®  the  mm-rt^f^rmtrre  wnj^Soitkm  obtainable  from  the  tube. 

Ai  a  nil*  moet  diftwlty  b  eip^iaaced  at  radio  frcsquencies.  Even  if  regeneration  is 
pmnrated  at  the  working  freqwney  c4  tfee  amplifier,  there  may  be  considerable  feedback 
^^1?****?:  tf  *  r*WM1*Ilt  ^^  e*i^  •*  «ae  <rf  these  high  frequencies  a  parasitic 
^^wc«»*«w  »aj  bepn.  This  may  ea&s©  the  amplifier  to  cease  functioning  as  an  ampli- 
Itar.  and  wro^  p^ectiw  ^vi«  limits  the  plat©  current  the  tubes  may  be  over- 

heated  wy  "^ 


REGENERATION  AND  ITS  PREVENTION 


7-29 


Another  type  of  parasitic  oscillation  may  exist  in  the  pushpull  circuit  shown  in  Fig.  31 ; 
in  which  the  input  circuit  has  a  certain  impedance  from  ground  to  grids,  because  of  the 
leakage  in  the  center-tapped  grid  and  plate  inductances.  This  condition  effectively  places 
the  pushpull  tubes  in  parallel  with  an  input  and  output  circuit  with  the  neutralizing  con- 
densers ineffective.  If  the  input  and  output  circuits  have  sufficiently  high  impedance 
(which  may  be  comparatively  low  for  high-power  tubes)  parasitic  oscillations  will  result. 
However,  it  should  be  noted  that  TI  and  r£  in  Fig.  31  are  connected  from  their  respective 
grids  to  ground  so  that  the  resistors  are  still  effective  in  suppressing  parallel  parasitic  oscil- 
lation of  the  tubes.  If  the  corresponding  value  of  resistance  in  r\  and  r2  is  placed  across 
the  grids  without  the  r-f  ground  at  the  center  point,  parallel  parasitic  oscillation  of  the 
tubes  may  exist  because  the  grids  would  be  at  essentially  the  same  potential  so  that  the 
resistances  would  be  ineffective. 

This  type  of  parasitic  oscillation  is  apparently  responsible  for  the  self-oscillation  of 
most  pushpull  audio  amplifiers. 

PREVENTION  OF  OSCILLATION.  Regeneration  to  the  point  of  oscillation  may  be 
prevented  in  several  ways.  It  is  often  accomplished  by  the  introduction  of  power-absorb- 
ing resistors  into  various  parts  of  the  amplifier  circuit,  but  these  invariably  increase  the 
losses  in  the  circuit  and  so  are  objectionable.  Another  possible  method  is  to  decrease 
the  impedance  of  the  input  circuit,  but  this  is  frequently  impracticable  without  too  great 
decrease  in  amplification. 

Parasitic  oscillations  may  sometimes  be  prevented  by  so  changing  either  the  plate  or 
grid  circuit  that  the  resonant  frequencies  of  the  one  have  no  counterpart  in  the  other. 
There  is  then  no  complete  resonant  path  at  any  frequency.  In  general,  if  the  leads  to  the 
grid  are  short,  and  a  small  r-f  choke  is  inserted  in  series  with  the  plate  to  increase  the  effec- 
tive length  of  the  plate  leads,  there  will  be  no  common  resonant  frequency.  If  the  circuits 
cannot  be  so  unbalanced  resistances  must  be  used,  such  as  are  shown  in  Fig.  31. 

The  most  effective  cure  for  parasitic  oscillation  of  pushpull  audio  amplifiers  is  to  un- 
balance the  input  and  output  circuits  by  placing  a  resistor  or  small  condenser  across  one 
side  of  the  input  or  output  transformers.  In  general,  the  value  of  impedance  required 
to  suppress  the  undesired  oscillation  will  not  noticeably  affect  the  frequency  characteristic 
of  the  amplifier. 

Improvement  in  shielding  between  stages  will  sometimes  eliminate  enough  of  the  feed- 
back to  make  the  circuit  stable,  or  the  plate  and  grid  circuits  may  be  moved  apart  if  prac- 
ticable. Shielding  for  pentode  or  tetrode  r-f  amplifiers  is  always  necessary.  Shielding  an 
audio  amplifier  is  usually  not  necessary  with  proper  placement  of  the  various  tubes  and 
transformers  for  minimum  feedback.  However,  it  may  be  necessary  to  use  elaborate 
means  to  prevent  stray  pick-up  to  the  input  transformer.  In  general,  a  magnetic  shield 
for  audio  frequencies  is  better  than  a  copper  shield,  but  the  copper  is  better  at  radio  fre- 
quencies. 

Where  the  feedback  is  due  to  the  grid-plate  capacitance,  this  capacitance  may  be 
eliminated  by  electrostatically  shielding  the  plate  from  the  grid.  This  has  been  done  in 
the  tetrode  and  pentode.  It  makes  possible  the  extremely  high  amplification  factors 
obtainable  with  these  tubes. 

NEUTRALIZATION.  When  it  is  impossible  to  eliminate  the  fed-back  voltage  it  is 
still  possible  to  cancel  its  effect  by  introducing  an  equal  and  opposite  voltage.  Circuits  to 
accomplish  this  are  based  on  the  impedance  bridge  principle  or  on  the  three-winding  trans- 
former. (See  p.  6-12.) 

The  first  method  for  neutralizing  the  grid-plate  capacitance  was  that  of  Rice  shown  in 
Fig.  36.  Diagram  A  of  this  figure  is  a  schematic  sketch  of  connections;  diagram  B  is  the 
resolution  of  the  circuit  into  that  of  the  three-winding  transformer.  From  the  theory  of 


Input 


rJ£-C^ 

6  }  *•  \c 

L2 

n 

^Ci 
.11 

[llflj 

7 

U 

C2 

II 

OlK 

u 

JIQQOOQOOO, 

^ 

,0    Output 

!LI 

It            a 

e 

L 

No  Voltage  If 
Unity  Couplinr 


:°3 


A.  Schematic  Diagram 


B.  Equivalent  Circuit 


FIG.  36.     Rice  System  for  Neutralization.    The  tuning  and  bypass  capacitors  shown  in  the  circuit 
diagram  on  the  left  are  omitted  from  the  equivalent  three-winding  transformer  diagram  on  the  right. 


7-30 


VACUUM-TUBE  CIRCUIT  ELEMENTS 


tlie  ttsree-wiiwiiag  transformer  any  voltage  induced  in  the  coil  a-c  will  cause  no  voltage  drop 
mmzm  b-0re  (except  that  due  to  incidental  resistance  of  the  coils)  if  the  value  of  On  is 
properly  ebcnea  and  tfac  coupling  between  6-a  and  a-e  is  unity.  In  practice  the  con- 
denser  (C«)  is  ma4e  variable  to  permit  its  proper  adjustment  and  is  called  the  neutralizing 
coiicleriiaer.  The  disadvantages  of  the  Rice  method  lie  in  the  fact  that  part  of  the  input 
voltage  acrws  ®~«  is  toot  since  it  is  not  impressed  on  the  grid,  and  that  neither  side  of  the 
nfrutralixin^  condenser  can  be  grounded. 

A  nwlificatkMi  of  the  Rice  method  is  shown  in  Fig.  37,  A  being  a  schematic  diagram 
©f  epaiiiscikKia.  Diagram  B  shows  that  this  circuit  can  also  be  resolved  into  a  three- 
wiodiiig  traawfornMT  circuit.  From  the  theory  there  will  be  a  value  of  C«.  for  which  the 
iroltagft  acn^is  6»«  wiH  be  very  k»w,  approaching  zero  as  the  resistance  of  c-a-f  decreases. 
This  system  is  not  so  useful  for  power  amplifiers  as  the  Rice  system;  the  plate  circuit  of 
ll»®  fetter  taa  be  more  favorably  loaded  because  the  entire  plate  coil  is  in  the  plate  circuit. 


Schematic  Diagram 


.  Equivalent  Circuit 


37. 


Modified  Eke  System  for  Neutralization.     The  tuning  and  bypass  capacitors  shown  in 
;  ctofcKram  <m  the  kit  are  outfitted  from  the  equivalent  three-winding  diagram  on  the  right. 

Both  th«ae  nfcfthodi  require  that  for  perfect  neutralization  the  coefficient  of  coupling 
be  unity  and  thafe  the  incidental  resistances  be  sero;  hence  perfect  neutralization  can  be 
only  spfsreaeiitfci. 

A  p^shpnil  r-l  amplifier  can  be  neutralized,  as  shown  in  Fig.  31,  by  means  of  the  con- 
dcnaera  C't  and  C»,  tlw  value  of  which  is  equal  to  the  grid-plate  capacitance  of  the  tubes 
for  symmetrical  input  and  output  circuits. 

In  general.  slikMiag  of  the  input  and  output  circuits  of  a  capacitance-neutralized 
ampJilser  a«»©d  M>$  be  so  complete  or  effective  to  prevent  feedback  as  for  the  screen-grid 
tube  amplifier,  foemaae  some  stray  magnetic  or  capacitance  feedback  can  be  effectively 
^©d  fw  fc$r  tfae  proper  adjustment  of  the  neutralising  condensers  for  minimum 
A  eoTOBEiQia  and  convenient  method  to  adjust  the  neutralising  capacitance 
.  i  to  apply  a  signal  to  the  grid  of  the  amplifier  to  be  neutralised  with  the  tube  in 
placw  ami  the  fil&m©i^l*ifet*sd  but  without  plate  voltage.  An  r-f  galvanometer  is  placed 
m  %b®  plat®  tank  ctrtoit,  asd  tlie  neutralising  capacitances  are  adjusted  for  zero  or  mini- 
mnm  galvaMMBetet  deHeetkm.  Precaution  against  exce^ive  currents  through  the  gal- 
TMwmrtw  must  be  taken  for  the  initial  adjustment  of  the  neutralizing  condensers,  or 
other  metb&cb  of  indkatiag  &  tank  voltage  may  be  used. 

If  the  amplifiwr  to  be  i&eutratiwed  is  a  plate-modulated  class  C  amplifier  the  simplest 
and  quickest  met&od  for  neutralising  is  to  overmodulate  the  amplifier  and  adjust  the 
neutraiiainK  «i»ndeiM^r  until  the  instantaneous  value  of  carrier  will  be  zero  at  the  peaks 
of  dowinpard  modulation.  A  cathode-ray  oscillograph  may  be  used  as  an  indicator  for 

BIBLIOGRAPHY 

timely  Small  Tubes,  Proc.  I.R.E.,  July,  1931. 

is  Class  B  and  Class  C  Amplifiers,  Proc.  I.B.J?.,  March, 

J£  X,  Gf*$$itai  DetenninAtkm  <^  Performanee  of  Push-poll  Audio  Amplifiers,  Proc.  I.R.B., 
A,  ?,  Ll«chnnf  Th«  Output  Characteristics  of  Amplifier  Tubes,  Proc.  I.R.B., 
^*Z^^^i  ^3^^  £ly  ^J^^l  referring  to  the  Proc.  I.R.E. 

«"-«- 


WIDE-BAND  AMPLIFIERS  7-31 

SPECIAL-PURPOSE  AMPLIFIERS 

By  E.  L.  Clark 

A  special-purpose  amplifier,  as  the  name  implies,  is  a  particular  type  of  vacuum-tube 
amplifier  designed  for  a  definite  purpose.  Such  amplifiers  are  usually  of  the  capacitance, 
impedance-coupled  circuit  type,  as  differentiated  from  tuned  and  mutually  coupled 
amplifiers. 

A  wide-band  amplifier  is  an  amplifier  that  covers  a  wide  frequency  spectrum,  such  as 
is  used  for  a  television  video  amplifier.  In  most  cases,  the  wide-band  amplifier  operates  as 
a  class  A  amplifier  with  plate  current  flowing  for  360  electrical  degrees.  A  wide-band 
amplifier  is  characterized  by  low  gain;  for  a  given  tube  the  gain  multiplied  by  the  band 
width  is  a  constant. 

A  cathode  follower  is  an  amplifier  which  has  its  output  load  in  the  cathode  circuit  of  a 
vacuum  tube.  The  characteristics  of  a  cathode  follower  are:  reduced  input  capacitance, 
increased  input  resistance,  reduced  output  impedance,  a  gain  of  less  than  1,  and  no  voltage 
inversion.  The  cathode  follower  may  be  used  as  an  impedance  matching  device,  without 
the  use  of  a  transformer. 

A  grounded-grid  amplifier  is  an  amplifier  that  has  its  grid  grounded,  and  the  input 
signal  applied  to  the  cathode.  The  output  circuit  is  in  the  plate,  in  the  usual  manner. 
The  characteristics  of  a  grounded-grid  amplifier  are:  low  input  impedance,  low  input-to- 
output  capacitance  due  to  the  shielding  action  of  the  grounded  grid,  and  no  voltage 
inversion. 

An  in-pnase  amplifier  is  one  hi  which  the  polarity  of  the  output  signal  is  the  same  as 
that  of  the  input  signal.  This  is  of  no  importance  in  sine-wave  work.  However,  for  pulse 
work  and  for  television  video  amplifiers,  the  polarity  of  the  amplified  signal  is  of  great 
importance. 

A  negative  feedback  amplifier  is  an  amplifier  in  which  some  of  the  output  signal  is 
fed  back  to  the  input  to  modify  the  output.  There  are  two  general  types  of  feedback 
amplifiers,  the  voltage-feedback  and  the  current-feedback  types.  Both  are  characterized 
by  a  reduction  in  gain.  However,  the  voltage-feedback  type  gives  an  apparent  decrease 
in  plate  resistance.  Both  types  of  feedback  give  a  reduction  in  distortion. 

A  one-shot  amplifier  is  an  amplifier  which,  after  responding  to  an  input  pulse,  will  not 
respond  to  a  second  pulse  until  a  given  time  has  elapsed.  This  amplifier  is  similar  to  a 
multivibrator,  and  it  requires  two  tubes.  The  characteristic  of  a  one-shot  amplifier  is 
that,  after  a  given  input  level  has  been  reached,  the  output  is  a  sudden  sharp  pulse  which 
cannot  be  immediately  repeated,  a  definite  time  interval  being  required  before  a  second 
pulse  can  be  obtained. 

A  pulse  amplifier  is  an  amplifier  that  is  designed  to  handle  a  pulse  type  of  input  signal. 
Pulse  signals  are  of  two  general  types:  positive  pulses  and  negative  pulses.  The  pulse 
amplifier  must  be  designed  for  the  polarity  of  pulse  to  obtain  the  optimum  performance. 
The  band  width  of  the  pulse  amplifier  must  also  be  adjusted  to  the  type  of  pulse  being 
handled  if  the  pulse  shape  is  not  to  be  degraded  and  if  maximum  gain  is  to  be  obtained. 

The  characteristic  of  a  pulse  amplifier  intended  to  handle  positive  pulses  requires  that 
it  be  biased  nearly  to  cutoff  and  that  it  shall  approach  class  B  operation  and  efficiency. 
However,  to  handle  negative  pulses  the  vacuum-tube  bias  must  be  near  zero,  and  a  heavy 
plate  current  is  drawn  except  when  the  pulse  signal  is  applied.  This  results  in  low  effi- 
ciency for  a  negative  pulse  amplifier. 

5.  WIDE-BAND  AMPLIFIERS 

The  wide-band  amplifier  is  used  when  the  frequency  response  must  be  extended  beyond 
about  15  kc.  It  finds  its  chief  application  as  a  video  amplifier  in  television  equipment. 
However,  it  is  finding  other  uses  such  as  in  radar  and  pulse  work.  In  order  better  to 
understand  the  frequency  response  limitations  of  an  r-c  coupled  amplifier,  see  Fig.  1. 
This  is  a  curve  of  output  voltage  vs.  frequency,  of  a  typical  amplifier  stage,  as  shown  in 
Fig.  2.  The  frequencies  5/i  and  h  are  considered  the  useful  limits  of  the  wide-band  am- 
plifier. This  is  somewhat  empirical,  but  practice  has  shown  it  to  be  desirable  in  order  to 
reduce  the  low-frequency  phase  shift  (see  p.  7-46).  If  many  cascaded  stages  are  used 
it  may  be  desirable  to  take  10/i  instead  of  5/i  as  the  low-frequency  limit  of  the  amplifier. 

HIGH-FREQUENCY  RESPONSE.  The  high-frequency  range  will  be  defined  as  that 
region  lying  above  about  1  kc.  The  simplest  form  of  wide-band  amplifier  is  that  shown 
in  Fig.  2  with  a  low  value  of  plate  resistor  (rz,).  In  general,  pentodes  are  used  for  wide- 


7-32 


VACUUM-TUBE  CIRCUIT  ELEMENTS 


band  amplifiers,  for  two  reasons.  First,  the  Miller  capacitance  effect  is  negligible;  second, 
they  are  made  with  a,  higher  mutual  conductance  (gm)  than  triodes.  The  general  formulas 
for  a  class  A  amplifier,  eqs.  (1),  (la»,  and  (2)  (p.  7-03),  also  cover  wide-band  amplifiers. 
However,  the  value  of  Z  is  of  interest  and  governs  the  gain  and  band  width  of  the  amplifier. 
In  Fig.  2*  Or  is  the  total  shunting  capacitance,  composed  of  the  stray  circuit  capacitance, 
the  raitput  capacitance  of  Fj,  and  the  input  capacitance  of  V$  (which  may  be  partly 
Miller  capacitance  in  triodes).  Let  the  resistance  r0  represent  the  resulting 


100 

I 

|70.7 


/.       A 


FIG.  I.     Ampli&er  Resixmse  Curve 
Cs 

Hh 


1! 


-*-+ 


-JL- 

*  +B       -C 

Fio.  2.     Uz^mipaosated  Amplifier  Stage 


pwalk!  r«i^iuaee  of  tfe*  |^lat€  reebtor  TL  ai^i  the  grkJ  resistor  rx.    The  absolute  value  of 
Z  a  given  by 


A*  th»  ^r^^»^  j^  wU^  imakes  t}^  ©aimcitance  reactance  **  e<jual  to  r0,  the  response 
inn  be  ckmn  to  7^.7  per  «»t  i^  the  maximum  response.  This  gives  a  means  of  determin- 
ing $h®  valu*  el  f  r- 


A*  tk9  plM«  r««^or  ri  in  «»m%  m«di  smalfer  thaa  rr?  the  effect  of  rg  is  usually  negligible 
IMC  !«.  »*»•  aratoal  oomtectance  of  Ube  vacuum  tube,  the  gain  of  the  stage  is 


Gam  A  = 


n  becomes 
in  .4  -  gmZ 


«reil*Br 


stage  is 

(3) 
',  therefore 

(4) 


(5) 


WIDE-BAND  AMPLIFIERS 


7-33 


for  parallel  resistance  and  capacitance  as  shown  in  Fig.  2  which  represents  a  time  delay  of 

0.035 


j 


•  seconds 


(6) 


For  an  un  compensated  amplifier:  Fig.  3  gives  the  value  of  TL  for  various  values  of  shunt 
capacitance  CT,  at  the  frequency  /2.  The  response  will  be  down  to  70.7  per  cent  of  the 
fiat  portion  of  the  curve  at  frequency  /« when  using  a  plate  load  TL  as  determined  from  Fig.  3. 


10,000. 


100. 


1,0 
Frequency  /2,  megacycles 


10.0 


FIG.  3.     Plate  Load  (TL}  in  Terms  of  Frequency  (/2>  and  Total  Capacitance  ((7jO  for  an  Uncempen- 
sated  or  Shunt  Peaked  Amplifier 

SHUNT  PEAKTTTG.  In  Fig.  4  is  shown  the  schematic  of  an  amplifier  stage  using 
shunt  peaking.  As  previously  described,  Ct  is  the  total  shunting  capacitance.  The  gain 
for  this  amplifier  is  given  by  eq.  (4) .  The  value  of  Z  is 

(7) 


+B          -C 
FIG.  4.     Circuit  for  a  Shunt  Peaked  Amplifier  Stage 

If  an  inductance  LI  is  chosen  so  that  XLI  =  J/2  %CT  at  /2,  the  resulting  impedance  Z  will  be 
equal  to  rz,.    This  means  that  the  response  will  be  flat  to  frequency  /2  instead  of  being 


7-34 


VACUUM-TUBE  CIRCUIT  ELEMENTS 


40C 


1000  2000  3000  4000 

Plate  load  rL,  ohms 


5000  6000 


5,    Shunt 


in  Terras  of  Frequency  f/j)  and  Plate  Load  (TL)  for  a  Shunt  Peaked 
Amplifier 


10,000. 


.  X     ,.    i    !x          X 


X  iX  i         X        \       x 


X  X  X  i      X      X 


X \i    i  \|  !  |.\     \!  \ 


\  i    \  I 


X    l\ 


\gj 


^xi 


x<? 


\ 


^     2a-/2CT" 


s 


&L 


\ 


\ 


\ 


\ 


\ 


\ 


1.0 

g,  megacycles 


\ 


\ 


\ 


\ 


\ 


\ 


\ 


\ 


\ 


10.0 
(Cr)  for  a  Shunt  Peaked 


WIDE-BAND  AMPLIFIERS 


7-35 


down  to  70.7  per  cent  as  in  the  uncompensated  amplifier.    The  values  are 

1 


(8) 


T  L 

Li  =  -— 

43T/2 


(9) 
(10) 


Gain  A  =  gmrL 

The  time  delay  is  no  greater  than  0.023//2  seconds. 

Figure  5  gives  the  value  of  inductance  LI,  for  various  frequencies  fa  and  load  resistance 
TL  for  use  in  a  shunt  peaked  amplifier. 

A  shunt  peaked  amplifier  can  be  designed  quickly  by  the  use  of  Fig.  3  and  Fig.  5.  Know- 
ing what  frequency  /2  is  needed  (say  3.0  megacycles),  determine  the  value  of  Ct\ 


400 


1000 


2000  3000  4000 

Plate  load  rt,  ohms 


5000 


6000 


FIG.  7.     Shunt  Inductance  (Li)  in  Terms  of  Frequency  (/2)  and  Plate  Load  Resistance  (TX)  for  a  Shunt 
Peaked  Amplifier  with  Corrected  Phase  and  Amplitude  Characteristics 

a  good  estimate.  Then  from  Fig.  3  find  TL  to  be  1330  ohms.  Now  from  Fig.  5  at  3  mega- 
cycles and  1330  ohms,  LI  is  found  to  be  35.2  microhenrys.  The  stage  gain  will  be  1330 
times  the  mutual  conductance  gm  of  the  tube  in  mhos. 

For  multistage  wide-band  amplifiers  where  the  best  phase  and  amplitude  character- 
istics are  needed,  slight  revisions  in  eqs.  (8)  and  (9),  as  shown  by  Freeman  and  Schantz, 
will  give  almost  perfect  results  up  to  frequency  /$ 


TL  = 


0.85 


0.353ri, 


(ID 


(12) 


Equations  (11)  and  (12)  are  plotted  in  Fig.  6  and  Fig.  7  to  facilitate  the  design  of  such  an 
amplifier. 

SERIES  PEAKING.  Figure  8  is  the  schematic  diagram  of  a  series  peaked  amplifier. 
It  can  be  seen  from  Fig.  8  that  the  capacitance  is  split  by  the  inductance  L±.  This  results 
in  the  vacuum  tube  Vi  working  into  a  smaller  capacitance  than  in  the  previous  case  of 
shunt  peaking,  with  the  results  that  more  gain  is  obtained.  For  proper  operation  the 
ratio  of  Ci/Cg  =  x/2.  If  Ci/Cg  —  2  the  plate  load  resistor  TL  must  be  put  on  the  other  side 
of  the  series  inductance  La.  The  rule  is  to  keep  the  plate  load  resistance  TL  on  the  low- 


7-36 


VACUUM-TUBE  CIRCUIT  ELEMENTS 
L2    CG 

VV 


-I- 


^-4- 


+8 


Fm.  ft.     Circuit  for  a  Series  Peaked  Amplifier  wh 
ssck?  01  th*?  series  inductance  £3.    The  value  of  TL  for  series  peaking  is  given  by 

(13) 


TL 
TL 


__ 

4rCi/s 

'fjert  <  V  Cj  "•*/!.    In  i&oet  ca^es*  however,  it  is  more  convenient  to  use  CT,  the  sum  of 
>  f  "j,  an<J  atrmya,  onee  C'r  ma  be  determined  more  accurately  than  Ci-    In  terms  of  Cr, 

j.  m  ci^^a  fey 


^ 

If  the  »»ri@§  iWbctj^©©  Z«  is  c&oaen  to  resonate  with  d  at  a  frequency  of  /s^a,  the 
witt  be  $a&  lo  f  rwineiicy  /s.    Tb©  yalne  of  the  ser^  inductance  L*  is  given  by 


%  m  tenm  of  C 

SiibsiJtwUFtiMc  £ 

10,000, 

y,  £3  »  © 

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10.0 
f^t  megacycles 

•ad  Total  Capacitanee  (Cf)  for  a  Series  Peaked 


WIDE-BAND  AMPLIFIERS 


7-37 


The  phase  delay  is  a  rather  complicated  function  in  the  series  peaking  circuit,  but  in 
general,  as  Seeley  and  Kim  ball  have  shown,  the  time  delay  up  to  the  frequency  ft  is  con- 
stant within  a  variation  of  0.0113//2  seconds.  This  is  roughly  one-half  the  variation  in 


800 


1000 


2000  3000  4000 

Plate  load  rt,  ohms 


5000 


6000 


FIG.  10.     Series  Inductance  (L2>  in  Terms  of  Frequency  (/2>  and  Plate  Load  (TX)  for  a  Series  Peaked 

Amplifier  Stage 

phase  delay  experienced  with  shunt  peaking.  The  gain  of  a  series  peaked  amplifier  is 
given  by  eq.  (10),  using  the  value  of  TL  as  obtained  from  eq.  (14). 

To  facilitate  the  design  of  a  series  peaked  amplifier  eq.  (14)  is  plotted  in  Fig.  9  and  eq. 
(16)  is  plotted  in  Fig.  10.  The  curves  of  Fig.  9  and  10  are  to  be  used  as  described  in  shunt 
peaking. 

COMBINATION  OF  SHUNT  AND  SERIES  PEAKING.  As  might  be  expected,  the 
advantages  of  shunt  and  series  peaking  can  be  combined  to  increase  the  gain  further. 


-C  +  +B  -C  + 

FIG.  11.     Circuit  for  a  Combined  Shunt  and  Series  Peaked  Amplifier  Stage 

The  series  inductance  La  separates  the  output  capacitance  Ci  and  the  input  capacitance 
Cs,  while  the  shunt  inductance  LI  compensates  for  the  output  capacitance  Ci.  The  cir- 
cuit is  shown  in  Fig.  11. 


?~38  VACTUM-TUBB  CIRCUIT  ELEMENTS 

10,000, 


rx 


I 


<g  1000. 

-2 

at 

£ 


s: 


X       TV 


^!~T 


N 


\ 


X 


N 


\ 


^-VJ 


^U 


\ 


>j  i 


j\ 


01 


ffl 

"~^\ 


X 


^ 


SJ 


\ 


N 


IX 


\ 


\ 


\ 


\ 


\ 


\ 


1.0 


Frequency  fzf  megacycles 


10.0 


F»«,  12.     H»*e  L©«d  CrU  ia  Tetms  ol  Freqweacy  (/2)  and  Total  Capacitance  (Cr)  for  a  Combined 
Shunt-serres  Peaked  Amplifier  Stage 


2000  3000  4000 

Plate  load  rt,ohms 


5000  6000 


WIDE-BAND   AMPLIFIERS 


7-39 


Again,  as  in  series  peaking,  the  ratio  of  Ci/C2  =  */2  for  proper  operation. 


1.8 


LI  =  0.12CFTZ,2 


If  the  value  of  CT  from  eq.  (17)  is  substituted  in  eq.  (18)  and  (19),  they  become 


0.936rL 


(17) 

(IS) 
(19) 


(20) 

(21) 


The  gain  of  a  combined  peaked  amplifier  is  given  by  eq.  (10)  using  the  value  of  TL  as 
obtained  from  eq.  (17). 


800 


5000 


6000 


°0  1000  2000  3000  4000 

Plate  Load  ru  ohms 

FIG.  14.     Series  Inductance  (£2)  in  Terms  of  Frequency  (f%)  and  Plate  Load  Resistance  (TL)  for  a 
Combined  Shunt-series  Peaked  Amplifier  Stage 

The  phase-delay  expression  becomes  more  complicated  in  this  type  of  peaking,  but  it 
does  not  exceed  0.015//2  seconds,  up  to  /2. 

If  the  Q  of  Z/2  is  too  high,  a  high-frequency  peak  will  be  experienced  just  before  f%  is 
reached.  The  resistance  r2  shunting  LZ  is  to  lower  the  Q  of  the  inductance  £2  and  prevent 
the  formation  of  a  peak  in  the  response  curve.  The  value  of  resistance  r^  may  vary  from 
5  to  10  times  the  load  resistance  TL. 

Equations  (17),  (20),  and  (21)  are  plotted  in  curves,  Figs.  12,  13,  and  14,  to  expedite 
the  design  of  a  combined  shunt-series  peaked  amplifier. 

CONSTANTS-TYPE  FILTER  COUPLING  NETWORK.  The  schematic  diagram, 
Fig.  15,  shows  a  wide-band  amplifier  of  the  constant-jK,  low-pass  filter  type.  This  circuit 
appears  at  first  glance  like  the  combined  shunt-series  peaking  network;  however,  its  con- 
stants are  based  on  standard  constant- JT  low-pass  filter  equations,  as  follows:  L%  =  — 
and  Cz  = In  forms  to  apply  to  the  circuit  of  Fig.  15  these  become: 

TL  =  -£gr  (22) 

Lz  =  ri?Cz  -  ^-  (23) 

la  -  ^  (24) 


7-40 


VACCUM-TOBE  CIRCUIT  ELEMENTS 


»n<I  r2  -  5  to  10  tin*  n.  The  stage  gain  is  still  given  by  eq.  (10)  iiang  the  valu of  r^ 
fr,«n  «*i.  «±J,  Thi-  Indicate  higher  gam  than  any  of  the  other  high  peaking  systems. 
Th»  li  true;  however,  one  factor  ha.  been  ignored  in  all  the  foregoing  systems,  namely 
capmciunce.  Only  the  distributed  capacitance  CA  across  the  series  coil  as 


Fro  13.     Circuit  fc*r  a  Wide-hand  Amplifier  with  a  Constant-JT  Configuration  Low-pass  Filter-coupling 

Network 

thown  IB  Fig,  15  h*a  mueh  effect.  This  capacitance,  however,  changes  the  seeming  con- 
*t*nt-K  tow-paw  filter  coupling  network  into  an  3£-derived  filter  section.  In  most  wide- 
bs«d  ampliSers  tb»  frequencies  encountered  are  high  enough  so  that  this  effect  cannot  be 
ignored.  In  changing  from  a  constant-lT  to  an  M -derived  low-pass  filter,  there  are  two 
ways  to  k«*p  a  giwn  pass  band:  to  reduce  the  shunting  capacitance,  or  to  reduce  the 
irapedimee.  As  the  capacitance  cannot  be  reduced,  the  only  course  is  to 

tbe  load  resistance  ri,  thus  lowering  the  gain.    The  equations,  taking  this  dis- 

I  capacitaacse  €4  into  consideration,  are  as  follows: 

'$  (25) 

(26) 

(27) 

fJS 

Also  Lj  «»  O.SZ-2  approximately,  and  r»  —  5  to  10  times  TL- 

Equmtioa  (28)  giv&s  the  frequcnc>*  of  infinite  attenuation  which  sliould  be  kept  well 
outaid*  tli@  pa^s  imad  to  preveat  excessive  phase  shift. 


M 


/* 


(28) 


Cnder  laorm&l  c*>aditioiis  the  phase  shift  is  about  the  same  as  for  the  combined  shunt 

ttd  series  peaking  ^nst«ni. 
Eqaatk>a  (25)  is  plotted  ia  Fig.  16,  which  gives  the  values  of  M  in  terms  of  the  output 

1.2  r 


1JJ 
O.s 


9,4 


as 

RfltlO  CEf  • 


Fs*.  ML    V*tat  si  Mm  Frodwed  bw 

a»d  the 


0.5 


C&p«^taiacc  Wd)  ol  tiie  Series  Peaking  Coil  ( 
»ri  (Rg.  15)  —  ^  %>«"  \ 


WIDE-BAND  AMPLIFIERS 


7-41 


capacitance  Ci  and  the  distributed  capacitance  C<t  of  the  series  inductance  L«.  The  curve 
in  Fig.  17  shows  the  effect  of  JJ  on  the  load  resistance  TL  and  the  series  inductance  L$.  As 
the  stage  gain  is  directly  proportional  to  the  load  resistance  TL,  the  importance  of  keeping 
the  distributed  capacitance  Cd  as  low  as  possible  is  apparent. 


120 


3-0  0.9  0.8  0.7  0.6  0.5  Q.4 

Value  of  M 

FIG.  17.     The  Effect  of  M  on  the  Plate  Load  (TL)  and  the  Series  Peaking  Inductance  I 


Equations  (22),  (23),  and  (24)  are  plotted  in  the  form  of  curves  in  Figs.  18  and  19  to  be 
used  for  designing  wide-band  amplifiers  of  the  constant-.??  configuration.  Knowing  the 
top  frequency  ft  (say  3  megacycles),  determine  the  value  of  C*;  25  /*/if  is  a  good  estimate. 
Then  from  Fig.  18  find  TL  to  be  4240  ohms.  Now  from  Fig.  19  at  3  megacycles  and  4240 
ohms  find  L$  =  448  microhenrys,  and  LI  =  224.  However,  the  distributed  capacitance 
of  Z/2  must  be  considered.  It  may  be  about  4  /z^f  while  Ci  may  be  approximately  16  jujuf. 


10,000. 


100. 


0.1 


1.0 

Frequency  /2,  megacycles 


10.0 


FIG.  18.     Plate  Load  Resistance  (rz,)  in  Terms  of  Frequency  (/a)  and  Input  Capacitance  (Ca)  for  a 
Constant-.^  Type  Low-pass  Coupling  Network 

This  gives  a  ratio  for  Cd/Ci  of  0.25;  referring  to  curve  Fig.  16  M  is  found  to  be  0.78.  Re- 
ferring to -Fig.  17  at  the  point  M  =  0.78  the  value  of  TL  will  be  78  per  cent  of  that  for  a 
constant-K  network  and  the  value  of  L$  will  be  60.8  per  cent  of  that  for  a  constant-J^ 
network.  These  factors  modify  the  value  obtained  above,  TL  becomes  3300  ohms,  and  £.3 


7-42 


VACUUM-TUBE  CIRCUIT  ELEMENTS 


900 


0 

F&s,  if 


IQOQ 


5000 


6000 


2QQQ  3GCQ  4000 

Bate  load  rv,ohms 

InduetMMM  (!i)  Bad  (La)  in  Terms  of  Hate  Load  Resistance  (rx)  and 
>  fur  a  Constant-^  Type  Low-pass  Filter-coupling  Network 

.„-  ,_i  272  izucrohe&ryB.    From  the  above  derivation  Li  =  218  microhenrys,  which  is 
appnmmattly  that  obtarod  from  Fig.  19.    The  shunting  resistor  r,  would  have  a  value 
•onMrfcm  fattwwn  15,000  cfens  and  33,000  ohms,  depending  upon  "the  Q  of  the  peaking 
Its  exact  TmJiie  would  have  to  be  determined  by  test. 


Type  #f  r»d«a  Swwu  SifMl  Used  to  li*t 


L 


_r 


»-ft-o. 
c-£»u 


1.0     je 


Loading  Resistor  r2  Too  Large 
Fio.    ^.    Curve    Shape   Obtained  When    the 
Loading  Resistor  (r2)  Is  Too  Large 


5   i 


With 


Type) 


Fw. 


loading  R?sZstoi-r2Too  Small 


.  Shape   Obtained    When    the 

Loading  Resistor  (rj)  Is  Too  Small 


Plate  Resistor  rLToo  Large 

No         FIG,   24     Curre    Shape   Obtaiued    When    the 
Plate  Load  Rector  (J-L)  Is  Too  Large 


WIDE-BAND  AMPLIFIERS 


7-43 


observed  across  a  very  low  plate  resistance,  possibly  100  ohms.  Figure  20  shows  the 
response  of  the  wide-band  amplifier  of  the  constant-^  configuration  when  all  its  com- 
ponents are  adjusted  properly.  Figures  21  to  28  inclusive  give  the  response  obtained 
with  various  components  that  have  incorrect  values. 


Plate  Resistor  rL  Too  Small 

FIG.  25.     Curve   Shape  Obtained   When   the 
Plate  Load  Resistor  (rx)  Is  Too  Small 

Js        JmitT 


Series  Psakfng  CoFI  L2  Too  Small 

FIG.  27.     Curve  Shape  Obtained  When   the 
Series  Peaking  Coil  (£5)  Is  Too  Small 


J        V 


Series  Peaking  Coil  L2  Too  Urge 

FIG.  26.     Curve   Shape  Obtained  When  the 
Series  Peaking  Coil  (La)  Is  Too  Large 


Shunt  Peaking  Coil  Lj  Incorrect, 
Too  Large  or  Too  Small  About  The  Same, 

FIG.  28.     Curve   Shape  Obtained   When  the 
Shunt  Peaking  Coil  (Li)  Is  Incorrect 


Multistage  wide-band  amplifiers  using  pentodes  or  beam  power  tubes  offer  no  particular 
difficulty.  However,  if  a  multistage  triode  amplifier  is  to  be  designed,  trouble  will  be  en- 
countered if  peaking  is  used  in  the  grid  circuit  and  plate  circuit  of  the  same  tube,  as  it 
may  result  in  a  tuned-grid,  tuned-plate  oscillator. 

FIGURE  OF  MERIT  FOR  WIDE-BAND  AMPLIFIER  TUBES.  The  capability  of  a 
vacuum  tube  to  amplify  at  high  frequencies  depends  not  only  upon  its  mutual  conduct- 
ance but  also  upon  its  input  and  output  capacitances.  It  is  these  capacitances,  in  addi- 
tion to  the  stray  capacitance,  that  limits  the  value  of  plate  resistance  TL  that  can  be  used. 
The  "Miller"  capacitance  effect  is  the  chief  reason  triodes  are  not  satisfactory  as  wide- 
band amplifiers. 

Considering  these  factors,  eq.  (29)  gives  an  acceptable  figure  of  merit  (F.M.)  for  wide- 
band amplifiers. 


F.M.  = 


gm 


Csk 


pk  -f  Cgp(l  +  JL) 


(29) 


Equation  (29)  is  applicable  to  either  triodes  or  pentodes;  for  pentodes  and  beam  power 
tubes  Cgp  is  so  small  that  the  last  term  may  be  ignored,  hence: 


F.M.  =  • 


(30) 


-f  Cant 
Table  1  gives  a  list  of  vacuum  tubes  with  their  corresponding  figure  of  merit. 

Table  1.    List  of  Vacuum  Tubes  Applicable  to  Wide-band  Amplifier  Service,  and  Their 

Figure  of  Merit 


Tube  Type 

Input 
Capacitance, 

Output 
Capacitance, 

Gm 

F.M. 

6AK5 
6AG5 

4.0 
6.5 

2.8 
1.8 

5,100 
5,000 

750 
602 

6AC7/I852 

11.0 

5.0 

9,000 

562 

6AG7 

13.0 

7.5 

11,000 

536 

6AU6 

5.5 

5.0 

5,200 

495 

6BA6 

5.5 

5.0 

4,400 

419 

6AB7/1853 

8.0 

5.0 

5,000 

385 

6SH7 

8.5 

7.0 

4,900 

316 

6SG7 

8.5 

7.0 

4,700 

303 

6L6 

10.  0 

12.0 

6,000 

273 

6V6GT 

9.5 

7.5 

4,100 

241 

954 

3.4 

3.0 

1,400 

219 

6K6GT 

5.5 

6.0 

2,300 

200 

COMPARISON  OF  HIGH-FREQUENCY  COMPENSATION  METHODS.  Table  2 
gives  the  essential  design  data  for  high-frequency  compensation  of  wide-band  amplifiers. 
The  last  three  types  listed  as  "practical  results"  give  data  obtained  by  measuring  the 


7-44 


VACUUM-TUBE   CIRCUIT  ELEMENTS 


con*tam-£  network  response,  with  different  ratios  of  d/C2.  As  the  distributed  capaci- 
tance <•  d  winnot  be  eliminated,  it  must  be  considered  as  producing  an  ^/-derived  low-pass 
filler  network. 

Table  3.    Summary  of  Wide-band  Amplifier  Formulas 


Type 


Relative 
Ll  Gain  at  /2 


M 


"C  wsora  p«?  i »&t  ®d 


0.707 


1.  00 


Sfaunt    f^*f   bwt 


0,85 


j     0.85 


Ci  'Ci  *  0.5  ,. 


1.5 


1.5 


1,8       i  0.52CJTL1     O.I2Crris       1-8 


Zr/jCr 


3.0 


C@«pl&!it  K  *nth  i 
Ci/Ci  *  S.5 


3.0M 


Practical  Results 


0.  8Z,2  a  pp. 


Crf 


OMMKUAI  K  wnfe  C< 


O.&Liapp. 


2.55Jf 


t  A'  »ith 
Ci/<  j  »  1.0 


O.SLs  app. 


^ 


Gain  «•  rx,Gjtf 

LOW-FEE  QtJEHCT  RESPGJTSB,    Low-frequency  atteauation  or  low-frequency  am- 
aad  plyise  ahift  may  be  introduced  in  any  one  of  four  places,  or  a  com- 
tbt  low.    Ttey  sare  (1  >  oilhode  re^stor  and  by-pass  condenser;  (2)  grid  con- 
tor  c^jplmg  ae^^trk;  (3)  sateen  aai^ly  re^stor  and  by-pass;  (4)  the  internal 

-  B  pow^r  supply. 

EPFBCT  OF  A  CA.THOBK  RESISTOR  A2TD  BY-PASS.  One  method  of  obtaining 
a  aeigftUY*  bm&  oa  tW  grid  of  a  vacuum-tube  amplifier  is  to  include  a  resistor  rk  in  series 
with  tJb*  o^toie  l»  ground;  to  preheat  low  of  gain,  this  resistor  r*  is  shunted  by  a  capaci- 
tor (V  S»  Fig-  39.  Th«  effect  of  this  Has  network  rjtC*  on  the  low-frequency  response 
by  tins  ffcet  tl**&,  the  feynrer  tlie  freqi^aey,  the  higher  the  capacitive  reactance  of 
the  k«i  ita  fllraatniff  effect  oa  r4.  This  results  in  cathode  degeneration  with  an 
loss  iri  (UA.  Tb«  gain  of  aneh  an  amplifier  stage  is  given  by: 


wad  the 


Gain  A  » 


(31) 


k  given  by 


C51  * 


ger.eral  fornraia.    For  a  wide-band  amplifier  the  tube  is  usually  of  the 
«>:»  O.  »»^  ^  »  1.0,  so  TLJT*  aad  !/*>  may  be  disregarded, 


<» 


WIDE-BAND  AMPLIFIEKS 


7-45 


To  prevent  loss  of  gain  at  the  lowest  frequency,  Zk  must  remain  essentially  constant.  In 
practice  this  may  mean  hundreds  of  microfarads  for  Ct,  especially  if  r*  is  low.  It  is  pos- 
sible to  compensate  for  the  loss  of  gain  due  to  cathode  degeneration  by  a  plate  filter 
in  the  plate  circuit  of  the  amplifier  stage  (Fig.  29).  The  conditions  that  must  be 


+B 


FIG.  29.     Amplifier  Stage  Considering  Low-frequency  Response  Only 

met  to  compensate  for  the  TkCk  network  are:  rtCj-  =  rpCp,  rp/rt  =  rigm,  and 
rLgm,  from  which  the  values  of  TF  and  Cp  are  obtained 


and 


(34) 
(35) 


THE  EFFECT  OF  THE  GRID  COUPLING  CAPACITOR-RESISTOR.  With  a  grid 
coupling  condenser  CG  and  resistor  rg,  Fig.  29,  the  voltage  63  impressed  on  the  grid  of  Fa 
will  decrease,  as  the  frequency  decreases,  assuming  e*  to  remain  constant.  The  ratio 
€3/62  is  given  by 


At  the  frequency  that  makes  the  capacitive  reactance  1/wCe  equal  to  the  grid  resistance 
r<y,  the  response  63/^2  will  be  70.7  per  cent  of  that  at  mid-range  frequency,  and  the  phase 
shift  will  be 

tan-i  if?  =  45  deg  (37) 

TG 

The  grid  resistor  TO  should  be  made  as  large  as  possible;  its  value,  however,  is  limited  by 
the  tube  manufacturer  to  a  maximum  for  a  given  tube  type.  For  a  given  low-frequency 
response  the  value  of  Co  may  have  to  be  so  large  that  there  is  danger  of  ruining  the  high- 
frequency  response  by  increased  stray  capacitance  to  ground.  In  practice  the  value  of 
CG  would  be  0.05  to  0.10  juf,  and  r<?  would  be  the  manufacturer's  maximum  value  for  the 
tube  type  being  used.  If  these  values  do  not  give  a  response  at  the  lowest  frequency  of 
5/i  as  indicated  on  Fig.  1T  compensation  is  needed, 

Equations  (36)  and  (37)  are  plotted  in  the  form  of  curves  in  Fig.  30.  This  gives  an 
easy  means  of  determining  low-frequency  response  before  compensation.  The  plate  filter 
CFTF,  Fig.  29,  can  be  so  proportioned  that  the  voltage  rise  across  TI£F  as  the  frequency 
decreases  can  just  compensate  for  the  voltage  loss  across  C<?,  thus  producing  fiat  response. 
Also  the  phase  angles  are  such  as  to  compensate.  To  achieve  this  compensation  the  time 
constants  of  the  plate-filter  circuit  and  grid  circuit  must  be  equal,  that  is,  r<?C<?  =  rx,CV, 
from  which 

CF  =  ^  (38) 


and 


(39) 


where  /  is  the  lowest  frequency  to  be  compensated  to  full  response.  These  equations  are 
based  on  the  fact  that  the  pentode  amplifier  is  a  constant  current  device  within  the  range 
of  operation.  The  value  of  rp  should  not  be  made  so  high  that  the  amplifier  plate  voltage 


7-46  VACUUM-TUBE  CIRCUIT  ELEMENTS 

fatt»  ai^pitwiablF  bekw  its  screen  voltage,  or  trouble  may  result  from  too  high  a  screen 


4iuftiip*tion. 

UtUA%  n  is  preferable  to  make  the  cathode  circuit  such  that  no  compensation  is  needed 
at  UM  kiwwi  frequency  required*  Then  compensate  the  grid  coupling  network  in  the 
pl»i«  circuit  by  m**am  of  the  plate  filter.  It  must  be  remembered  that  the  plate  filter 
coropeoMttion  will  tata  care  of  only  the  loss  of  lows  at  one  point.  Do  not  try  to  compensate 
for  f  if*  faHPtt  by  €  **£*£&  <s»m^«^i<m;  it  cannot  be  done. 


Phase  Angle,  degrees 


10  100  '  1000 

Frequency,  cycles  per  Second 

Fi»   30,     Ci*nre,  Giiing  tbt  Low-freqwaey  Respoiase  Obtained,  and  the  Low-frequency  Phase  Shift, 
wifcb  Different  (roCG)  Grid  Coupling  Network 

TH1  EFFECT  OF  THE  SCREEN  BY-PASS.  The  effect  of  n£z  (Fig.  29)  is  similar 
to  that  whidi  requite  from  cathode  degeneration  mentioned  previously.  However,  the 
ttreea  nirrfsfit  is  only  about  10  per  cent  of  the  plate  current,  and  the  screen-plate  mutual 
MMMluetA&c*  m  only  aboiat  12  per  cent  of  tlie  control  grid  or  cathode-to-plate  mutual  con- 
d^rtan  w.  and  m  the  effect  is  much  smaller. 

H  tb»  tniM»  aai^mBt  €^  r^a  ii  mad®  at  legist  4  times  as  long  as  the  period  of  the  lowest 
fraqtMmcy  it  is  dteaned  4a  pa«s,  tfee  efect  will  be  negligible.  The  value  of  r2  is  determined 
by  UM  nokac*  rtfQttirw&eiats  fe^*  th«  particiilar  tube  being  used;  then  the  value  of  C2  is 

lbj: 

Cs=5^  (40) 

THE  EFFECT  OF  tlttE  IWTIRHJLL  IMPED AlfCE  OF  THE  POWER  SUPPLY.     The 
supply  iBt«armal  ittp«diUM«  Zi  b  e^eatially  t^  reactance  of  the  output  filter  con- 
.    Tfcit  r«M9iMM9  bws>tin«^  common  for  all  ampHner  stages  and  may  result  in  the 
ol  bw«4r«»epi©ttey  nwponM  or  in  k>w-lre<|i*ency  motor-boating,  depending  upon  the 

if  $&»gt8,  tht  rt«pM^,  mud  the  gam  of  the  system. 

It  h^  been  f<atsd  that  tfee  empaciuve  remetaac«  ore  of  the  final  filter  condenser  in  the 
twm  MHPply  «>^*®m,  »t  tb«  bwwt  frequency-  encountered,  should  be  no  greater  than  10  per 
_„  of  tiw  ^««4iw  fmi  load  r«sstan»  to  prei-ent  (^mrnon  coupling  through  the  B  supply. 
Tl&t  vmtiift  ^  tfe«  fiaal  filter  capacitance  is  then  given  by: 


~- 

2w/r 
r  t«  th*  *fft4*i<*t  k«d  mwUnee  which  is  given  by: 


B  c«rr«at  of  the  B  supply. 


(41) 


(42) 


CATHODE  FOLLOWERS 


7-47 


In  general,  when  designed  for  low-frequency  response,  the  cathode  circuit  is  by-passed 
with  a  capacitor  of  sufficient  size  to  prevent  cathode  degeneration  at  the  lowest  frequency 
encountered,  or  the  cathode  is  grounded  and  negative  bias  is  supplied  to  the  grid.  The 
screen  circuit  is  adequately  by-passed  so  that  no  loss  of  lows  results,  and  the  B  supply 
impedance  is  made  sufficiently  low  to  prevent  trouble.  This  leaves  only  the  effect  of  the 
grid  capacitor-resistor  network  which  must  be  compensated.  This  compensation  is  done 
in  the  plate  circuit  of  tube  Vi  driving  the  grid  of  "Fa,  as  previously  explained. 


6.  CATHODE  FOLLOWERS 

The  name  "cathode  follower"  is  given  to  an  amplifier  stage  when  the  load,  or  the  major 
portion  of  the  load,  is  in  the  cathode  circuit  instead  of  in  the  plate  circuit.  Figure  31  shows 
such  an  amplifier  stage. 

This  type  of  operation  is  called  a  cathode  follower  because  the  cathode  tends  to  follow 
the  grid  in  voltage  as  signal  is  applied,  thus  reducing  the  actual  grid  to  cathode  voltage 

Piste 


Output 


Cathode 


Output 

PIG.  31.     Cathode-follower  Stage  General  Case, 

•with    Resistance   in    Both    the   Plate    and    the 

Cathode  Circuits 


Output 


FIG.    32.     Cathode-follower    Arranged 

for  Proper  Bias  by  Returning  the  Grid 

Resistor   to   a   Tap    on   the   Cathode 

Resistor 


below  that  of  the  applied  signal.  This  type  of  circuit  finds  its  widest  use  as  an  impedance- 
changing  device.  It  is  used  extensively  in  connection  with  wide-band  amplifiers  to  match 
a  line  where  a  transformer  would  be  impractical. 

Cathode  followers  usually  use  triodes,  since  high  gain  cannot  be  realized  and  since  the 
shunt  capacitance  of  a  pentode  is  high,  including  screen-to-plate  and  screen  by-pass 
capacitance  as  well  as  the  usual  cathode  capacitance. 

Cathode  followers  have  many  characteristics  not  found  in  amplifiers  of  other  types: 
(1)  output  from  the  cathode  circuit,  (2)  voltage  gain  to  the  cathode  of  less  than  1,  (3)  re- 
duction of  input  capacitance,  (4)  increase  in  input  resistance,  (5)  low  output  impedance, 
(6)  increase  in  effective  plate  impedance,  (7)  no  change  in  polarity  of  output  signal,  and 
(8)  use  as  a  phase  splitter  with  load  in  both  cathode  and  plate. 

When  a  tube  is  operated  as  a  cathode  follower,  the  circuit  should  be  so  arranged  as  to 
supply  the  proper  negative  bias  for  the  type  of  tube  being  used.  If  the  circuit  is  as  shown 
in  Fig.  31,  the  IT  drop  across  the  cathode  resistor  Tk  should  be  such  as  to  provide  the  proper 
bias  for  class  A  operation.  When  a  high  value  of  cathode  resistor  is  needed,  the  arrange- 
ment shown  in  Fig.  32  should  be  used.  The  IT  drop  across  the  resistor  TI  should  provide 
the  proper  negative  bias  for  class  A  operation.  If  the  cathode  resistor  is  so  low  in  value 
that  sufficient  negative  bias  is  not  developed,  an  external  negative  voltage  must  be  sup- 
plied to  the  grid  to  produce  proper  class  A  operation. 

In  the  general  case  (Fig.  31)  the  gain  to  the  cathode  is  always  less  than  1  and  is: 


Gain  A  = 

If  the  plate  resistor  is  zero  (Fig.  32),  eq.  (43)  becomes 
Gain  A 


(43) 


§H_± (44) 

1  +  rrfU/rj,)  +  gm] 

When  the  cathode  load  is  not  a  pure  resistance,  r*  is  replaced  by  Z*,  which  is  the  absolute 
value  of  impedance  at  the  frequency  of  interest. 


7*48  VACUUM-TUBE  CIRCUIT  ELEMENTS 

For  pentock  operfttxm  i>  y>  r&  »  1,  and  so  the  gain  to  the  cathode  becomes 

Gain  A  -  ,  f"r*  (45) 

1  4-  rfcgw 

The  effecthre  input  cmpadtaac©  of  a  cathode  follower  is  given  by 

cf, 


When  the  plate  resistor  ii  aero,,  eq.  (46)  becom.es: 

C-  -  C*  ('  -  l+nK!£)+,J  +  C"  (47) 

The  effective  input  resistance  is  given  by: 

r«H  -  -  ^  -  (48) 


-    __   m     _ 

1  4-  (ri/Tf)  +  rjfc[(l/rp)  4-  &*] 
Tfcf  effeniT^  ofitptit  impedance  r§  of  a  cathode  follower,  for  the  general  case,  is  given  by: 

-1 

(49) 


(49)  ia  for  the  t^be  alone  and  do&s  not  include  the  effect  of  the  cathode  resistor 
is  in  afaunt  with  rt,    Tlie  resulting  impedance  -Zii  is  given  by: 


the  plate  nwador  n  is  aero,  eq.  (49)  becomes: 

1 


(50) 


(51) 


In  a  pentode  r»  »  I,  and  so  r§  «  1  'gw. 

In  some  fa*ea  it  may  be  moc«  desirable  to  have  Zi  the  effective  output  impedance  of  a 
fftthodf  follower  ia  a  wngle  equation  rather  than  first  to  calculate  rfl  and  then  JZi.  The 
general  ernwe  for  Zi  directly  is 

Z     «sr  rl.  4"  Tp 

3  *  1  4-  r^»  4-  (1/r*)]  4-  (rx/rjb}  (52) 

Wheis  TI  fa  awto,  e*Q.  {521  b©c©i»^: 

y  1 

*«  **  /•.  j_  .    .   .,  >    >.    , (53) 

to  couple  into  a  line,  matching  its  char- 
tfeis  ^m  be  done  with  a  matching  output 

a        .     ,    ...  -apliSer  wf»-k  sadi  a  transformer  is  not  readily  available 

A  vtttnnae  faii@iwer  i^s  fe^  Q^^  i^.  g^^j  service. 

^ "^*.?%e?^*  ^r1*11"  ^  °<  ^«  Hoe  to  be  matched  is  less  than  r0  for  the 


th«j ^TMterMe  i»p«d«ioe  Z  of  tfee  line  is  hi^ber  than  the 

*    ±^*^  "*** "  "^ (Re-  M)-  *"  vai»e  ^^ tbe 

(~       \  {  gmfa+Z)  \ 

\r,  4-  2/  \1  4-  (r,  -f  Z)^  4-  (l/r,)]/  (55> 


GROUNDED-GEID  AMPLIFIER 


7-49 


the  tube  plate  current;  otherwise  the  cathode  follower  will  not  operate.  If  Tk  is  too  high, 
however,  the  operation  may  be  improved  by  terminating  the  line  of  Fig.  33  for  direct 
current  as  well  as  for  alternating  current. 

When  an  amplifier  operates  with  an  unbypassed  cathode  resistor  (a  type  of  cathode 
follower) ,  the  effective  plate  resistance  rp  is  increased. 

rp'  -  rP(l  -f  gmrk)  (56) 

When  a  cathode  follower  is  operated  with  a  resistor  in  the  plate  circuit  equal  to  the 
resistor  in  the  cathode  circuit,  it  is  termed  a  "phase  splitter."  That  is,  the  voltage  de- 
veloped in  the  plate  circuit  will  be  equal  to  the  voltage  developed  in  the  cathode  circuit. 


FIG.  33.  Cathode-follower  Circuit  Used 
to  Match  a  Transmission  Line,  When 
the  Characteristic  Impedance  (Z)  of  the 
Line  Is  Lower  Than  the  Cathode  Imped- 
ance (ro)  of  the  Tube  (Vi) 


FIG.  34.     Cathode-follower  Circuit  Used  to  Match 

a   Transmission    Line,    When    the    Characteristic 

Impedance  (Z)   of  the  Line  Is  Higher  than  the 

Cathode  Impedance  (ro)  of  the  Tube  (Yi) 


However,  the  polarity  of  the  voltage  in  the  plate  circuit  will  be  the  inverse  of  the  voltage 
in  the  cathode  circuit.  This  type  of  circuit  may  be  used  to  obtain  pushpull  operation 
from  a  single  amplifier.  The  gain  to  the  plate  will  be  given  by  eq.  (31),  and  the  gain  to 
the  cathode  will  be  given  by  eq.  (43). 


7.  GROTJNDED-GRTD  AMPLIFIER 


A  grounded-grid  amplifier  is  usually  a  triode  which  has  its  grid  grounded  and  the  input 
connected  to  the  cathode.  The  output  circuit  is  in  the  plate  in  the  usual  manner.  Fig- 
ure 35  shows  such  a  circuit.  A  triode  connected  in  this  manner  has  some  of  the  character- 
istics of  a  screen-grid  tube,  as  the  grid  acts  as  a  shield  between  the  input  and  output  cir- 
cuits. It  also  has  some  of  the  characteristics  of  a  cathode  follower,  as  the  cathode-input 
impedance  is  equivalent  to  that  of  a  cathode  follower  with  a  plate  load.  Another  feature 
of  the  cathode-input  amplifier  is  that  there  is  no 

voltage  inversion  between  the  input  signal  and  yt  ^L  1   Output 

the  output  signal  as  with  conventional  grid  input.  ' 

The  input  impedance  Z\  is  given  by:  t 

i  == 


1  +  rp[gm  +  (1/rjb)!  + 

The  input  impedance  Zi  of  a  cathode-input 
amplifier  may  be  adjusted  to  match  a  line  the 
same  as  a  cathode  follower.  Solving  eq.  (52)  for 
n  gives 


J/2  [1  +  (TL/rP)}  -  [gm  +  (l/r,)3 


(57)      •=- 
FIG. 


•f-B 


35. 


Circuit     of     the     Grounded-grid 

Amplifier 

The  effective  impedance  Zk  in  the  cathode  cir- 

cuit must  be  determined  before  it  is  possible  to  obtain  the  gain  to  the  plate  and  the 
effective  plate  resistance  rpf.  Referring  to  Fig.  35,  r»  represents  the  internal  impedance 
of  the  signal  input  source,  and  rt  is  the  cathode  resistor. 


4- 


(58) 


Where  a  transmission  line  is  being  matched,  rj  would  be  replaced  by  Z,  the  character- 
istic impedance  of  the  line. 


7-50  VACUUM-TUBE  CIRCUIT  ELEMENTS 

The  fun  of  the  cathode-input  amplifier  is  given  by  e*.  (3D  using  Z*  as  obtained  from 
Gain  A  ~ ^-~ = '  <31> 


It  <an  be  SWB  from  eq.  (31)  that  to  obtain  high  gain  with  a  cathode-input  amplifier  the 
aubode  impedance  Z*  should  be  kept  low  and  TL  should  be  made  as  high  as  possible. 
Making  ft  high  «bo  increases  the  input  impedance  as  shown  by  eq.  (52). 

Tin*  effectiw  plmte  impedance  r/  of  a  cathode-input  amplifier  is  given  by  eq.  (56), 
•ubetttutiac  ZA  as  obtained  from  eq.  (58) 

r/  -  rp(l  +  j?raZ4)  (56a) 

wtiieh  giv*s  as  increase  in  plate  resistance  over  that  of  the  tube  operated  in  the  conven- 
tUMml  manner* 

a  m-PHASE  AMPLIFIERS 

The  name  in-phaae  amplifier  applies  to  an  amplifier  that  has  the  same  polarity  of  signal 
in  the  output  aa  thai  applied  to  the  input.  In  sine-wave  operation  this  is  of  little  impor- 
tanr«<  but  for  puisw  amplifiers,  or  television  amplifiers,  the  polarity  of  the  signal  is  impor- 
tant. Ttese  types  of  signals  are  not  symmetrical  about  an  a-c  axis  and  must  be  treated 


Thm  are  four  ®&H;eml  typee  of  in-phase  amplifiers: 

1.  Catted®  follower. 

2.  Oth<xi*-input  ampliEer. 

3.  Combined  cathode  follower,  cathode-input  amplifier  (cathode-coupled  amplifier). 

4.  Suppressor  input,  screen  output  amplifier. 

Tb©  cathode  follower  and  cathode-input-type  amplifiers  have  been  covered  in  the 
lMV**dutg  aertkNMi  and  will  not  be  discussed  further, 

The  third  typt  of  in-pim»e  amplifier  is  a  combination  of  cathode  follower  and  cathode- 
iiipiit  am plifer,  termed  cathode-coupled  amplifier.  This  type  of  amplifier  is  best  made 
by  nan*  a  dual  trk*$e  with  a  common  cathode,  such  as  a  6J6  or  a  6SN7  tube.  The  circuit 

is  shown  in  Fig.  36.    In  an  ampli- 
-**       fier  of  this  *ype  the  polarity  of 
utpu       the  signal  is  unchanged  at  any 
point  through  the  stage. 

The  value  of  rt  should  be  such 
that  the  negative  bias  developed 
by  the  combined  plate  currents 
of  FI  and  FS  is  proper  for  class  A 
operation  of  the  tubes  FI  and  Fa. 
In  operation  the  cathode-coupled 
amplifier  exhibits  characteristics 
F***»  m.  Th»  Ia-pfe»w  Ampiiiw  of  tbe  Catbode-coupied  Type  *&  some  respects  similar  to  those 

of  a  screen-grid  tube.    It  can  be 

*»cl  wkli  tuiied  input  &ad  tuued  outjmt  circuita  without  danger  of  oscillation,  such  as 
would  omir  if  a  trk*de  in  conventional  circuits  were  used. 

Tlfee  gam  ®£  (V\)  the  cathode  follower  can  be  calculated  by  using  eq.  (31),  the  value  of 
Zi  obtaffied  from  ©q.  (52 j  being  mibstituted  for  Z*.  These  two  equations  can  be  com- 
bai»d  to  give  the  gam  to  the  cathode  of  FI;  assuming  that  both  tubes  are  the  same,  then 

Gain  to  cathode  of  FI  = gm^L  +  r^ (59) 

The  c&thode  feHower  is  workiag  into  the  cathode  of  the  second  tube  which  exhibits  a 
tow  ca&!»de4npm  impcdbuce,  as  does  any  cathode-input  amplifier.  This  cathode  im- 
ptdaiw  must  be  comstt&mi  in  the  gain  equation.  This  is  done  by  using  Z±  of  the  cathode- 
istSWf  tube  aa  th»  rathodae  lo&d  for  tb.«  psi.tKfvl»  fnHmcaT-  Tk^  ;0  «  Tn*^.A.  ^-^T..-  J.T J.-L..J. 


.  ±  - 

liabe  M  the  cmtboae  toad  for  tfee  catliocte  follower.    This  is  a  lower  value  than  that 


. 

Im  obtammjc  the  gum  of  the  cathode-input  section  F2,  the  cathode-output  impedance 
for  the  catliodt  follower  Vj  must  be  used  for  the  cathode  load  of  the  cathode-input 
nplit^r  F^  TEe  Talw  of  Zi  from  eq.  (53)  must  be  substituted  for  Zt  in  eq.  (31).  Com- 
Ifa^we  two  equati0im,  the  gman  of  the  cathode  input  section  is  given  by: 


Oafa  fawn  «^b©a«  to  p48*e  of  V,  -  -  j^l  -|-  (ry/r^)  +  rrfm] 

2(1  -f  r^w)  +  (?L  +  rp)/rk  +  rm 


NEGATIVE-FEEDBACK  AMPLIFIEES  7-51 

The  overall  gain  of  the  cathode-coupled  amplifier  is  given  by  the  product  of  the  gain  to 
the  cathode  of  Vi  times  the  gain  from  the  cathode  to  the  plate  of  \\.  This  can  be  written 
in  a  single  equation  as 

Overall  gain  of  cathode-coupled  stage 

=  _  g^TL(TL  +  rp)[l  -f  (rp/ra)  -f  rpgm]  _ 
J2(l  +  rpgm)  +  [<rL  +  rpj  /rfc]  -f  ri,[(l/r,)  -f  gm}\^ 

SUPPRESSOR  INPUT,  SCREEN  OUTPUT  AMPLIFIER,    The  circuit  for  this  ampli- 
fier is  given  in  Fig.  37.    The  circuit  requirements  are 

as  follows  :  the  tube  should  be  of  a  type  that  normally  j  *""*"        v 

operates  with  the  screen  at  a  lower  voltage  than  the 
plate,  and  also  it  must  have  a  suppressor  grid  struc- 
ture  that  has  an  effective  mutual  conductance  to  the 
plate.  The  6AS6  is  a  satisfactory  tube  for  this  type  of 
operation,  as  the  suppressor-grid-to-plate  has  a  mu- 
tual conductance  of  about  1000  micromhos,  and  the 
suppressor  grid-to-screen-grid  has  a  mutual  conduct- 
ance gms  of  about  850  micromhos.  The  screen  im- 
pedance Z,  is  of  the  order  of  10,000  ohms. 

By  the  use  of  a  6AS6  for  an  in-phase  amplifier,  the 
values  shown  in  Fig.  37  give  satisfactory  results. 
Care  must  be  taken  when  operating  a  tube  in  this 
manner  not  to  exceed  the  screen  dissipation. 

The  gain  A  to  the  screen  is  given  by  eq.  (3)  using  r,  in  place  of  Z  and  the  screen  im- 
pedance Za  in  place  of  rp;  the  equation  then  becomes 


Gain  A  =  (62) 

Tt  -J-  ^* 

where  gms  is  the  mutual  conductance  of  the  suppressor  10  the  screen. 

9.  NEGATIVE-FEEDBACK  AMPLIFIERS 

In  the  negative-feedback  amplifier  a  voltage  obtained  from  the  amplifier  output  is  fed 
back  to  the  input  in  such  a  way  as  to  oppose  the  applied  signal. 

There  are  two  general  types  of  negative  feedback.  The  first  is  negative  voltage  feed- 
back which  occurs  when  a  fraction  /3  of  the  output  proportional  to  the  voltage  across  the 
output  load  is  fed  back  to  the  input.  The  second  is  negative  current  feedback,  which 
occurs  when  the  voltage  fed  back  is  proportional  to  the  current  through  the  output  load. 
The  major  difference  in  the  results  produced  by  voltage  and  current  feedback  are  that 
negative  voltage  feedback  results  in  a  reduction  of  the  effective  internal  resistance  of  the 
amplifier,  whereas  negative  current  feedback  produces  an  effective  increase  in  the  internal 
resistance  of  the  amplifier. 

Owing  to  the  effect  of  reactance  in  the  circuit,  the  voltage  which  is  fed  back  may  not 
be  wholly  out  of  phase  with  the  input  voltage.  This  phase  shift  is  likely  to  occur  at  ex- 
tremely low  or  extremely  high  frequencies.  In  a  single  stage  it  cannot  exceed  90°T  which 
results  in  no  feedback  and  a  corresponding  increase  in  gain.  When  more  than  one  stage 
is  included  in  the  feedback  amplifier,  the  phase  shift  may  exceed  90  °,  which  results  in 
regeneration  and  perhaps  oscillation.  The  method  used  to  combat  this  phase  shift  and 
resulting  oscillation  is  to  make  one  stage  with  a  narrower  band  width  than  the  others. 
This  should  result  in  a  loss  of  gain  through  the  narrow  stage  to  a  value  at  which  oscilla- 
tion cannot  occur,  by  the  time  the  phase  shift  has  exceeded  90°. 

The  effects  of  negative  voltage  feedback  are  (1)  reduction  in  gain,  (2)  reduction  in  dis- 
tortion, (3)  reduction  in  noise,  (4)  improvement  in  the  fidelity  with  frequency,  (5)  greater 
consistency  of  characteristics  with  changes  in  applied  voltages,  and  (6)  reduction  of  the 
effective  internal  resistance  of  the  final  amplifier  stage. 

The  gain  or  amplification  in  the  presence  of  voltage  feedback  is  given  by  the  relation 

Gain  with  feedback  A'  =  ,    A  ...  (63) 

1  —  Ap 

However,  for  negative  feedback  $  is  negative  and  the  relations  become 

Gain  with  negative  feedback  A'  =  a  (64) 

1  -f-  j$,p 

which  is  a  reduction  in  gain  from  A,  the  amplification  without  feedback. 


7-52 


YACUUM-TUBE  CIKCUIT  EUEMENTS 


When  the  value  of  A  w  large  in  comparison  to  1,  the  gain  becomes  practically  inde- 
pendent of  the  amplifier  characteristics,  becoming  approximately  A'  —  l/£. 

Negative  voltage  feedback  reduces  the  non-linear  harmonic  distortion  produced  in  the 
amplifier  for  a  given  output  voltage  according  to  the  relation 

(IX)  Distortion  with  negative  feedback  -  R  (65) 

1  -p  Ap 

wikere  JE>  is  ti*e  distortion  with  no  feedback.  This  assumes  that  no  distortion  is  produced 
wb«  reamplifving  the  distortion  voltages  fed  back,  which  is  quite  accurate  if  the  distor- 
tM»  with  no  feedback  is  not  large. 

Feedback  will  reduce  the  distortion  up  to  a  certain  point,  but  feedback  cannot  increase 
the  power-output  capabilities  of  a  given  amplifier.  The  distortion  will  be  low  over  a 
portion  of  the  output  range  but  then  will  increase  faster  than  for  an  amplifier  with  no 
feedback  (Fig,.  38). 


Relative  Power  Output 
Fio,  38.    The  Effect  of  Negative  Voltage  Feedback  on  Distortion 

Negative  voltage  feedback  will  improve  the  signal-to-noise  ratio,  if  the  source  of  noise 
k  m  the  amplifier  and  not  fed  in  as  part  of  the  input  signal.    For  this  case,  assuming  equal 


Signal  to  noise  with  feedback     _          A1 
Signal  to  noise  without  feedback  ~  Az(l  -f-  Ap) 


(66) 


4j  is  tlie  amplification  from  the  point  of  introduction  of  the  noise  to  the  output 
with  negativfc  feedback,  and  AI  the  amplification  from  the  point  of  noise  introduction  to 
tte  cmtput  wHIioQt  feedback. 

With  »£g»f  tv*  feedback,  as  tli*  response  starts  to  drop  with  either  high  or  low  frequency, 
the  £e«dfeaefc  also  decreases,  opposing  the  change  and  resulting  in  a  flatter  frequency 

***MB***s**aM1'     Tfcus 

(67) 


Gam  at/2<^t)       ^4S{1 

f )  feedback  causes  an  apparent  reduction  in  the  plate  resistance  of  the 
.  em  a  feedback  amplifier.    Hie  actual  plate  resistance  rp  does  not  change 

j  aay  impeduM  mataunc  (as  with  an  output  transformer)  should  be  done  on  the  basis' 
of  DO  feedback.  Ifcen.  with  feed^ck,  the  response  and  damping  effect  as  on  a  loud- 
9^t®T  TWOQAOM  is  the  mm®  m  though  the  plate  resistance  rp  were  lowered  to  r  '  The 
dfotiv*  pl*U  r^^we«  r/  is  i|»  same  whether  the  feedback  is  over  a  single  stage  or 
,  provided  (he  g»ia  reduction  is  the  same, 
i  by 


I  +  (i/rf)  -  r+T^  (68) 

s  with  negative  voltage  feedback  applied  to  a  single 


NEGATIVE-FEEDBACK  AMPLIFIERS 


7-53 


The  gain  of  such  an  amplifier,  A',  taking  feedback  into  consideration,  is  given  by 

Af  =  1     A.  (69) 

1  —  Aa 

but,  since  a.  has  a  negative  sign  for  negative  feedback,  eq.  (69)  in  reality  should  be 

A'  ~  rri^  <70) 

where  a.  is  the  ratio  of  the  feedback  resistor  rn  to  the  load  resistance  TL,  and  A  is  the  am- 
plifier gain  without  feedback. 


FIG   39.     Typical  Negative  Voltage  Feedback  Circuits  Applied  to  a  Single  Stage  of  Amplification 


The  effective  plate  resistance  rp'  is  given  by  the  expression 

TJ  =  rp(l  +  gmrn) 


(71) 


where  gm  is  the  mutual  conductance  of  the  amplifier  output  stage. 

The  increase  of  input  resistance  is  similar  to  that  obtained  with  a  cathode  follower, 
Equation  (48)  will  give  the  effective  input  resistance;  Tk  should  be  replaced  with  rn. 

The  input  capacitance  is  given  by  eq.  (46)  by  replacing  r*  with  rn.    Equations  (46)  and 
(47)  apply  to  current  feedback  over  a  single  stage. 
.    Figure  40  shows  some  typical  circuits  with  negative  current  feedback. 

THE  OKE-SHOT  AMPLIFIER.  The  one-shot  amplifier  is  a  form  of  multivibrator 
with  one  of  the  tubes  biased  beyond  cutoff.  The  circuit  is  shown  in  Fig.  41  .  Under  steady- 
state  conditions  tube  Vi  is  cut  off  and  tube  V+  draws  plate  current.  The  circuit  remains 
in  this  condition  until  a  positive  trigger  voltage  of  sufficient  amplitude  to  cause  Vi  to 


T-54 


VACUUM-TUBE  CIBCUIT  ELEMENTS 


Fw.  40.     Tjrpiml  Nf«»4iT*  Cmrreat  Feedback  Circuite 


current 

charging  the  coupling  ca- 
pacitances Ci  and  C2,  result- 
ing in  a  large  negative  po- 
tential on  the  grid  of  FI. 
As  soon  as  the  current  in  the 
plate  circuit  of  Vi  stops  in- 
creasing, the  voltage  on  the 
grid  of  F2  starts  to  rise;  this 
is  also  regenerative,  and  Fa 
resumes  its  steady-state 
plate  current,  but  Vi  is  left 
with  its  grid  highly  negative. 
This  negative  voltage  dis- 
charges through  rj.  exponen- 
tially back  to  the  applied 
bias  voltage  J&,  Fig.  42. 

Owing  to  the  regenerative 
action  the  output  of  this 
type  of  amplifier  is  a  sharp 
pulse  of  short  duration.  The 
plate  of  Fi  gives  a  negative 
pulse,  and  the  output  from 
Fz  is  a  positive  pulse. 

The  voltages  acting  on 
the  grid  of  Fi  are  shown  in 
Fig,  42.  The  trigger  pulses 
should  be  limited  in  ampli- 
tude so  that,  at  the  desired 
time  after  firing,  the  ampli- 
fier is  again  ready  to  fire. 
"With  the  trigger  pulse 
limited  to  amplitude  P  it 
cannot  fire  the  one-shot  am- 
plifier earlier  than  the  pre- 
scribed time.  Assuming 
that  Ci  and  Ca  are  small 
enough  to  permit  them  to 
change  to  the  peak  positive 
voltage  applied,  then  the 
operating  conditions  can  be 
calculated. 


Ir  «irop  tbrauiift  the  plate  load  resistor  n,  of  Fa  must  be  appreciably  greater  than 
cutoff  voifakg*  r«qiwe4  cm  IV 


Positive 


.f  B  Negative  Pulse 

Circuit  Dtagram  of  ft  One-shot  AmpiiSer 


I*.  41  far 
totow^toU^cuw^b^riOt 


tiae  trigger  poise  amplitude  P 
bias  to  be  equal  to  ^i  -f  P/2, 


PULSE  AMPLIFIER 


7-55 


Zero  Bias 


HI 


FIG  42.     The  Voltage  on  the  Grid  of  Tube  (TO  of  Fig.  41  during  Operation 

The  time  T  that  must  elapse  between  firing  of  the  amplifier  and  the  second  firing  by  a 
pulse  of  amplitude  P  is  T  =  r^Ci  -f-  C2).  The  applied  bias  —  Ec  to  Vi  of  Fig.  41  is 
given  by: 

and  EB-L  is  given  by: 

BBI  =  Mitel*  -  1-85P)  (73) 

where  jui  is  the  amplification  constant  of  V\,  and  r^/p,  the  voltage  drop  across  the  plate  load 
rz,  of  FA,  is  assumed  to  be  the  maximum  negative  voltage  on  the  grid  of  "V\  immediately 
after  firing. 

10.  PULSE  AMPLIFIER 

With  the  advent  of  radar  and  television,  pulse  amplifiers  became  necessary.  The  pulse 
amplifier  is  an  adaptation  of  the  wide-band  amplifier.  The  bandwidth  necessary  is  de- 
pendent upon  the  pulse,  the  high-frequency  response  is  governed  by  the  rate  of  rise  and 
decay  of  the  pulset  and  the  low-frequency  response  is  determined  by  the  duration  of  the 
pulse.  (See  Section  9.) 

The  equivalent  frequency  /2  of  the  pulse  is  considered  to  be  equal  to  that  of  a  sine  wave 
that  rises  from  zero  voltage  to  peak  voltage  in  the  same  time  as  that  of  the  pulse.  This 
results  in  a  frequency  /2  that  must  be  passed  by  the  pulse  amplifier,  given  by 

/.  -  -4  w 


where  T  is  the  rise  time  or  decay  time  of  the  pulse,  whichever  is  shorter  (see  Fig.  43). 


T^  u-7*-*! 

H-TF       TH  H 


Symmetrical  Unsymmetrlcal 

Negative  Pulses 


FIG.  43.     Negative  and  Positive  Pulses,  Showing  the  Pulse  Rise  Time  (  Tr)  ,  the  Pulse  Decay  Time  (2»  , 

and  the  Pulse  Duration  (Tz>) 

Determine  the  frequency  /a  represented  by  the  rise  time,  then  refer  to  the  section  on 
wide-band  amplifiers  and  determine  the  design  of  an  amplifier  having  the  required  high- 


7-56 


VACUUM-TUBE  CIRCUIT  ELEMENTS 


-E. 


_l/p<ots*  input 


*H 

(a)  ^®p»r*i*nt  potat  for 


•Ovtpvt 


FWL 


44.  Grid!  Bias,  Plate  Curreat  Curve, 
im«  tfee  Operating  Point  £«<•  *  Positive 
,  tto  Opvnrtu*  Ptatn*  fw  a  Ne&mtiTfc  PuJsse, 
a  Cirroit  Th*t  Witt  AutMftftticaUy  Set  the 
»l  tibe  ProfNPt  Voltaot  for  Correct  Opera- 
®|  th*  Fyalw  Polarity  or  Ampli- 
tude 


frequency  response.  The  low-frequency  re- 
sponse at  5/i  needed  is  obtained  from  the  pulse 
duration  Tz>- 

fl  =  JL  (75) 

Again  refer  to  the  section  on  wide-band 
amplifiers  to  determine  the  constants  needed 
to  produce  the  required  response. 

The  consideration  for  the  initial  bias  point 
for  the  vacuum-tube  grid  is  somewhat  dif- 
ferent in  pulse  amplifiers  from  what  it  is  in 
conventional  amplifiers.  For  a  positive  pulse 
a  high  bias  is  needed.  This  bias  can  be, 
and  usually  is,  supplied  by  grid  current  (Fig. 
44,  Ai).  '  _  . 

For  a  negative  pulse  a  low  bias  is  needed 
and  the  tube  draws  a  heavy  current  except  in 
the  presence  of  the  pulse  which  drives  the 
tube  grid  toward  cutoff.  This  is  shown  in 
Fig.  44,  As. 

The  circuit  shown  in  Fig.  44,  J5,  is  ade- 
quate for  either  a  positive  or  a  negative  pulse 
amplifier.  The  time  constant  of  the  grid 
circuit  rjCi  should  be  several  times  the  pulse 
repetition  rate  to  maintain  bias  between 
pulses. 

With  positive  pulses  the  grid  is  biased  by 
means  of  grid-leak  bias  to  a  value  represented 
by  the  a-c  axis  of  the  positive  pulse.  This  is 
self-adjusting  and  requires  no  controls.  When 
one  is  operating  with  negative  pulses,  the  bias 
will  be  essentially  zero,  depending  upon  the 
pulse  a-c  axis,  and  will  operate  equally  as 
well  as  for  a  positive  pulse.  A  circuit  of  this 
type  is  self-adjusting  and  assumes  an  operat- 
ing point  so  as  to  provide  efficient  operation 
regardless  of  pulse  amplitude  or  polarity. 


^  (I-JF)  AMPLIFIERS 

By  Charles  J.  Hirscfc 

to  amplify  and  separate  signals  at  high  radio  frequencies,  some 
the  signal  frequency  to  a  fixed  intermediate 


it  m 
(kucwii  &i 

Ttif>  Mcn*l  tfcMMH  ia  mmtpliSed  arid  selected  at  the  new  frequency  by  means  of  an 
i~/  oMjRlv$*r.  Ssadb  rwwtwrs  differ  from  tuned-radio-frequency  (t-r-f)  receivers,  which 
amplify  by  maaas  of  eibreiiit®  tun&d  to  the  high  carrier  frequency. 

Th*  mtenoMNltttto  frw$u&eney  is  usually  lower  than  the  radio  frequency  and  higher  than 
tlwi  tnega*ncy  of  utiBsaAkm  (audio  or  video  frequency). 

Th*  »ieriiM»Ai»4«»-lr^qp@iic^'  ampliiser  has  the  function  of  amplifying  the  signals  within  a 
ipmifd  s-f  band  m®&  erf  rej^ctia^  all  others.    It  is  the  most  important  factor  in  the  deter- 
of  •encttivitjr.  sefectivity,  and  fi<felity  of  superheterod>Tie  receivers.    Since  these 
e  eonapl®^  r«K^iv€i*  are,  ia  the  main  part,  tite  characteristics  of  the  i-f 
to  maie  th*rn  ooDstam  over  the  tuning  range. 


UL  PACTORS  JJPfBCTHIG  THE  CHOICE  OF  INTERMEDIATE 

FREQUENCY 

Tl»  A«^w  of  imtem*«diat«  fnquency  rec^res  a  careful  study  of  the  following  factors:  (1) 
weraH  gMft,  (&  wlMtivity,  C3?  i»^e  rej^ctMm.,  (4)  tuning  range,  (5)  tweets  (whistles  caused 
by  faurvMHUNMol  tb*  i-4  «H»db  am  a»n*r»t*d  %  Hit  sec€»d<^eetorand  reimpressed  on  ther-f 
wnn-1*  to  b«ai  with  tb»  ««Aml  frequenej*),  (6)  i-f  re|©ctiois,  (7)  strong  stations  separated  by 
,  (8)  ©oa,  i^.,  number  ol  t«iw>d  <arcwits  and  their  components. 


FACTORS  AFFECTING  INTERMEDIATE   FREQUENCY      7-57 


Low  intermediate  frequencies  have  the  advantages  of  (a)  high  stage  gain  because  a 
higher  impedance  can  be  presented  to  the  output  of  the  amplifier  tube;  (b)  narrow  band 
width,  i.e.,  better  selectivity  because  a  given  frequency  separation  is  a  greater  fraction  of 
a  low  intermediate  frequency  than  of  a  high  intermediate  frequency  (see  Universal  selec- 
tivity curve,  Section  6) ;  and  (c)  greater  freedom  from  tweets  (when  the  i-f  amplifier  is  pre- 
ceded by  a  high  degree  of  r-f  selectivity)  because  only  higher  and  therefore  weaker  (but 
more  numerous)  harmonics  of  the  i-f  occur  at  the  signal  frequency  and  beat  with  the 
signal  to  produce  tweets. 

High  intermediate  frequencies  have  the  advantages  of  (a)  higher  irnage  rejection  by 
(1)  increasing  the  separation  (twice  intermediate  frequency)  between  the  desired  signal 
frequency  and  the  image  frequency,  and  (2)  reducing  or  even  eliminating  that  part  of  the 
tuning  range  within  which  signals  can  produce  images;  (6)  reduction  in  the  number  of 
"tweets"  because  fewer  harmonics  of  the  intermediate  frequency  lie  within  the  tuning 
range;  (c)  greater  freedom  from  "birdies"  (whistles  produced  by  combinations  of  r-f  sig- 
nals) because  combinations  of  r-f  signals,  separated  in  frequencies  by  the  intermediate 
or  subharmonics  of  the  intermediate  frequency,  will  not  be  impressed  on  the  converter  to 
produce  intermediate  frequency,  or  beat  with  the  local  oscillator  to  produce  intermediate 
frequency;  (d)  greater  freedom  of  interaction  (pulling)  between  the  local  oscillator  and 
the  antenna  circuit  because  of  greater  frequency  separation. 

High  image  rejection  and  freedom  from  "birdies"  require  costly  r-f  selectivity.  There- 
fore, a  high  intermediate  frequency  is  economical  because  it  reduces  the  requirements  for 
r-f  selectivity. 

The  i-f  amplifier  frequency  must  not  be  too  close  to  the  tuning  band  as  the  receiver  will 
then  become  unstable. 

Table  1  presents  a  comparison  of  the  receiver  characteristics  associated  with  two  inter- 
mediate frequencies  for  the  broadcast  band;  Table  2  gives  some  idea  of  the  intermediate 
frequencies  commonly  associated  with  specific  radio  frequencies. 

Table  1.     Comparison  of  Two  Radio  Receivers  Having  (a)  an  Intermediate  Frequency 
of  175  kc,  (6)  an  Intermediate  Frequency  of  455  kc 


Tuning  Range  550-1720  kc 


i-f 
I75kc 


i-f 
455  kc 


1.  Frequency    separation   between    desired 
station  and  image 

2.  Frequency  range  in  which  stations  within 
the  tuning  range  can  cause  images 

From 

To 

(Note:  The  ability  of  stations  outside  the 
tuning  range  to  produce  images  must  not 
be  overlooked.) 

3.  Frequency  range  of  stations  which  may 
be  interfered  with  by  images  produced  by 
stations  in  the  tuning  range.    (See  above.) 

From 

To 

4.  Harmonics  of  the  intermediate  frequency 
occurring  in  the  tuning  range 

5.  Separation  of  stations  capable  of  beating 
with  each  other  in  the  first  detector  to 
produce  intermediate  frequency 


2  X  175  =  350  kc 


550 -}- 2  X  175  -  900  kc 
I720kc 


550  kc 
1720  -  2  X  175  =  1370  kc 

4th,  5th,  6th,  7th,  8th,  9th 
175kc 


2  X  455  -  910  kc 


550  4-  2  X  455  =  1460  kc 
1720kc 


550  kc 
1720  -  2  X  455  =  810  kc 

2nd,  3rd 
455  kc 


Table  2.    Examples  of  Usual  Intermediate  Frequencies 


Tuning  Range,  Me 


(a)  0.150-0.275. 
(6)   0.150-L720. 


(c)    0.540-23.0. 


(d)  40-50  Me  F-m 

(e)  88-108  Me  F-m 

CO    54-88  and  174-2 16  Telev 

(g)   200  Me  pulse,  communication, 
(fc)    1000  Me  and  up 


Intermediate  Frequency,  Me 


0.130 

0.455-0.465 

with  some  European  receivers  at  0.260—0.360  to 
gain  more  selectivity 

0,455 

Some  receivers  use  0.455  Me  for  the  whole  fre- 
quency coverage.     Others  switch  to  an  i-f  of  2 
Me  when  receiving  signals  above  9  Me. 
4-5 
10.7 
20-30 

(21.75  sound — 26.25  picture) 
H. 7-15-30 
30-60  Me 


f-58 


VACUUM-TUB!!  CIRCUIT  ELEMENTS 


1*.  NARROW-  AND  MEDIUM-BANDWIDTH  I-F  AMPLIFIERS 

I-f  wnpKfim  am  be  etasiified  into  (a)  narrow  (10  kc)  and  medium  «  200  kc)  band- 
width, md  (&l  wkto-bftndwidtJi  (>  300  kc)  amplifiers.  .  .  . 

Hmrrew-  aad  medium-bandwidth  i-*  amplifiers,  which  are  used  in  most  receivers  receiving 
audio  f  iwmraty  »-m  m  f-m  isitettipmee,  usually  consist  of  individual  stages  using  pairs  of 
cotipfed-tuned  cimaita.  A  rwtanguiarty  shaped  selectivity  curve  (one  having  a  flat  top 

ami  ataap  «3®*t  usually  i*  desirable-  

AMPLIFIERS  FOR  A-M  BRQABCAST  RECEIVERS.    These  usually  consist  of 
&*  critically  omipW,  <k>yble-i*med  stages  at  a  frequency  between  455  and  465  kc. 


4250  v 


IVyfeal  14  Amplifier  for  iaeipensive  550-1720  kc  Receiver 


is  affected  by  the  need  for  (1)  gain,  (2)  selectivity,  (3)  fidelity,  (4)  stability, 
{&)  vDoutaHny. 

Gala  aad  S^^^trl^.  Tbe  i-C  aaapH^r  k  tfee  major  source  of  gain  and  selectivity  in 
&  radio  nmaivftr.  The  l-f  span  (froiu  radio  frequency  on  the  converter  grid  to  the  second 
dtotoctor)  dbpeffA  eta  tho  perfonxiaaee  required.  It  will  lie  between  a  rmmnnnm  of  1400 
for  «fj®4m^dl  (aaualbr  SS6~1750  kc)  sets  having  a  high  gain  r-f  stage  and  large  r-f  pick-up 
tip  to  &  auudmtmi  ®£  50,000  lor  rtwrt-wmv«  sets  with  small  r-f  pick-up  and  using  low-noise 
«3»?wt#r  tubia.  For  ffiiagb-foaiid  sets  it  is  seldom  necessary  to  use  more  than  one  i-f  stage 
I©  yet  the  r«*niir@d  gam  aad  sei^Jtivity.  In  general  in  such  cases  (two  i-f  transformers,  see 
F%.  1)  tba  ^a»e  «a«®  nomfe  be  M|£i.  TMa  reciuire®  high-impedance  transformers  which  in 
Mtra  l»plii«s  !$%ii  -t/C  r»t^  aisd  hlgji  Q  circuits  to  gain  the  required  selectivity  (see  eqs. 


Tva^aiii  I-f  Amplifier  U 


+1CX3  w  -1-250  v 
^x  Tmsed  Circuits 


«tM»). 

(pb 


TU  Q  m  saad*  ludi  by  using  (a)  lit*  wire,  (5)  iron-core  coils,  (c) 

A,  aud  (ili  large  shield  cans. 

tnti  gain  are  required,  and  in  general  for  multiband  sets,  two 
^dxtoeddrcuiHseeFig.^.    Tte  gain  and  selectivity  of 
e^  (I^^IS)  befew.    Values  for  typical  sets  are  sliowm  in  TaHe  3. 


NARROW-  AND  MEDIUM-BANDWIDTH  I-F  AMPLIFIERS      7-59 


Table  3, 


Average  Stage  Gains  and  Second  Detector  Sensitivities  for  Different  Types 
of  Broadcast  Receivers  Produced  between  1934  and  1946 


^^^^^              Gain 
Type  of  Set  ^^\^^ 

Conversion 
at  600  kc 
v 

1st  i-f 
V- 

2nd  i-f 
V 

Overall 
P 

2nd  Detector  Sensitivity 
Will  Produce 

i-f  Volts 
30%  Mod. 

a-f  Watts 
in  Voice  Cofl 

30 
23 
43 
32 
39 
40 
26 
14 
44 
39 

35 

10 
37 

94 
61 
88 
44 
56 
100 
56 
41 
120 
90 

2,800 
1,400 
2,800 
50,000 
2,200 
4,000 
14,300 
21,000 
5,300 
3,500 

0.3 
0.3 
0.5 
0.5 
0.5 
0.6-0.9 
0.6-0.9 
0.6-0.9 
0.7-1.1 
0.7-1.1 

0.05 
0.05 
0.05 
0.05 
0.05 
0.50 
0.50 
0.50 
0.50 
0.50 

Ac-dc  (r-f  gsin  of  6) 

Battery  (no  r-f  stage)    

(no  r~f  stage) 

(r-f  gain  of  10) 

Ac  (no  r-f  stage) 

(no  r-f  stage) 

(r-f  gain  of  1  2) 

Auto  (no  r-f)       

(r-f  gain  of  40)  

If  two  i-f  stages  are  used,  the  overall  gain  can  be  held  down  to  reasonable  values,  stabil- 
ity can  be  improved,  and  the  cost  can  be  reduced  by  using  (a)  solid  wire  coils  instead  of 
litz  as  the  five  or  six  tuned  circuits  will  supply  adequate  selectivity  even  with  the  lower  Q 
of  the  solid  wire  coils  (6)  lower  L/C  ratio,  (c)  output  voltage  obtained  from  a  tap  on  the 
secondary  (but  not  in  the  stage  feeding  the  diode,  as  a  condenser  across  the  diode  load  is 
necessary  to  present  a  low  reactance  to  i-f  harmonics  generated  by  the  diode),  (d)  unby- 
passed  cathode  resistor,  (e)  increased  bias  (however,  this  decreases  the  effect  of  avc  on 
this  tube). 

Gain  and  Selectivity  of  Last  Stage.  The  gain  of  the  stage  feeding  the  avc  diode  must 
be  high  enough  so  that  its  own  grid  will  not  overload  owing  to  inadequate  gain  control  of 
the  preceding  stage.  In  other  words,  it  must,  without  overloading,  supply  enough  power 
to  the  diode  so  that  the  diode  may  supply  adequate  control  voltage  to  the  preceding  stages 
for  all  signal  amplitudes  to  be  expected.  For  remote  cutoff  tubes  30  to  40  volts  of  avc 
may  well  be  needed,  which  represents  from  60  to  85  per  cent  of  the  peak  carrier  voltage 
impressed  on  the  diode. 

The  ratio  of  a-c  to  d-c  diode  load  impedance  should  be  as  near  unity  as  possible  to 
prevent  amplitude  distortion  on  high  percentage  modulation  signals. 

Variable  Selectivity.  (See  reference  4.)  Extreme  selectivity  usually  causes  extreme 
cutting  of  the  side  bands  with  loss  of  fidelity.  To  overcome  this,  some  receivers  use  varia- 
ble selectivity.  This  usually  is  obtained  by  coupled  circuits  which  are  critically  coupled 
(or  undercoupled)  when  selectivity  is  required,  but  which  are  overcoupled  when  selectivity 
can  be  sacrificed.  As  the  coupling  increases,  the  selectivity  of  each  coupled  pair  assumes 
the  well-known  two-peak  form.  The  frequency  separation  between  peaks  CA  —  /a)  in- 
creases with  the  coupling. —  =  K  approximately  for  overcoupled  high-£  circuits. 


Modulator 


K  Is  adjustable  frtwn 
K^Kcto  K>Kc 


Tl 


Low  M  to  approximate 

two  sfngfe-tuned 
circuits  in  cascade 
13  ^-^  T4 


FIG.  3.     I-f  Amplifier  Having  Variable  Selectivity 

The  valley  between  the  peaks  is  filled  in  by  the  selectivity  of  a  single^fcuned  circuit.  A 
flat-topped  selectivity  curve  can  be  approximated  for  all  coefficients  of  coupling  if  the  Q 
of  each  circuit  of  the  coupled  pair  is  equal  to  twice  the  Q  of  the  single-tuned  circuit.  In 
practice,  the  i-f  amplifier  takes  the  form  of  Fig.  3t  which  consists  of  three  double-tuned 
circuits.  The  coupling  of  the  first  two  pairs  is  adjustable  and  capable  of  being  overcoupled. 


7-60 


TACUUH-TUBE  CIBCUIT  ELEMENTS 


The  waging  of  the  teat  pair  k  low  so  as  to  approximate  the  selectivity  of  two  single-tuned 
circuit*  m  eaueade  Cos*  to  M  in  the  valky  of  each  overcoupled  pair).  The  Q  of  the  adjust- 
able pain  to  equal  to  twiee  the  Q  of  the  very  loosely  coupled  pairs.  The  cathode  resistors 
of  shout  im  ohma  ar*  unbypaaeed  to  minimize  the  detuning  caused  by  variation  of  the 
fetus  by  the  arc,  {3e#  reference  3.) 

BrtMbd  i-f  ampttfierB  are  sofnetimes  used  to  reduce  the  need  for  frequency  stability  in 
the  local  asettl&tor  in  pushbutton  aeta  or  short-wave  sets. 

Ttoatag  Stm&Ilty.  The  amplifier should  be  tuned  with  enough  capacitance  (>  25  yujuf)  so 
thai  it  will  not  be  appwtaMy  detuned  by  (a)  replacement  of  tubes,  (6)  displacement  of 
p*rte  by  Tifaratioa,  (r)  efaaag*  in  input  capacitance  of  vacuum  tubes  caused  by  variation  of 
gun.  CTtii  emu  fe«  ImiMsced  fey  a  Sinai  tinbypassed  cathode  resistor  (reference  3)  of  about 
HX>  ohms;  ssw?  Fi«. 


Bamiwlrftha 


Bami  width  at  6  db 
coupling  to  be  slightly 
below  optimum 


^^ 
~~2*7  '-1"' 


p«  la  t«  b«  !M»|  »  mtobnwii  by  3*pa*»ti*»«  plate  &  grfd  leads 


Me  14  Transformer 

r  will  rawi  wkk  Taiiatkjns  in  temperature  and  humidity,  the  use  of 
stable  ix€»d  uow^aaers  »ad  tmiBf  by  meiuis  oi  adjustabie  iron-core  coils  is  desirable. 

The  is*dy  itiw  ooupliac  few  ween  wmdiags  caa  either  aM  or  oppose  the  coupling  due  to 
the  eap«cii«a<*  fe«twe«a  the  pl»t«  and  gr»i  terminals  of  the  i-f  fe-ansformer. 

II  vety  «a^  traiwlormers  M«  dewred,  opfxieing  inductive  and  capacitive  coupling 
ptroits  ciowr  apaan«  of  the  winding.    F<r  coaxial  coils,  wound  in  the  same  direction 
permit  the  o«k  to  be  partly  self-skidded  eiectrostaticany  by  usin^ 
as  th#  fow  i-f  poteatial  terminals.  ^^ 

ia  th«s  ate^c  mp»eitaiice,  due  to  differences  in  production  wiring 
®ttw®®11  w^®8  *»  to  vitedik»,  causes  much  larger  variations  in  the 
"  ^  SSUe  with  ^^J^0^1^  «^l^ags  than  with  aiding  coupling 
quanfeities  variee  much  ^^r6  thaa  their  sum 


ccm^ing  AoiiW  be  sufficiently  below 
fc  «•  prr^wt  ww««»^te«  due  to  production  variations  and  vibration. 


NARROW-  AND  MEDIUM-BANDWIDTH  I-F  AMPLIFIERS      7-61 


With  either  coupling,  production  will  be  much  more  uniform  if  the  eapacitive  coupling 
is  made  as  small  as  possible  (see  Fig.  4). 

Feedback  of  I-f  Harmonics.  The  second  detector  output  must  be  well  filtered  and 
kept  as  far  as  possible  from  the  r-f  components  so  as  to  prevent  harmonics  of  the  inter- 
mediate frequency  from  being  impressed  on  the  r-f  circuits  where  they  can  beat  with  the 
signal  and  cause  "tweets." 

I-F  AMPLIFIERS  FOR  FM  RECEIVERS.  The  same  considerations  hold  for  fre- 
quency modulation  as  for  amplitude  modulation  except  that: 

1.  Bandwidth  must  accommodate  maximum  signal  frequency  swing  so  as  not  to  cause 
amplitude  distortion. 

2.  Gain  must  be  adequate  to  operate  limiter  on  weakest  signal. 

3.  Top  should  be  reasonably  flat  so  as  not  to  overtax  limiters  and  to  reduce  distortion 
in  balanced  detectors. 

3a.  Bandwidth  at  least       Table  4.    Typical  Dual  I-f  Amplifier  Stage  for  Am-Fm  Set 
150  kc  at  -6  db.     Adja- 
cent channel  attenuation 
±400  kc  at  least  50  db. 

4.  Selectivity    curve 
should  be  symmetrical. 

5.  Intermediate    fre- 
quency   for    the    88-108 
band  is  usually  10.7  Me. 

6.  Amplifier   consists 
usually  of  sets  of  double- 
tuned  circuits. 

7.  A-m  and  f-m  circuits 
usually  are  combined  in 
one    common   shield   can 

(but  care  must  be  taken  to        Note:  The  ratio  of  the  bandwidths  of  two  signals  20  db  and  6  db 
prevent  interaction  SO    stronger  than  the  signal  at  resonance  is  a  measure  of  the  coupling, 
that  leads  will  not  change    ™s  is  indicated  as  W^fW,  and  is  equal  to  2^37  when  LiCi  -  £2C2, 
v          ,          j  j-  Qi  ==  <?2,  and  the  circuits  are  critically  coupled.    When  Qi  ^  Q*  as  in 

couplings   by  adding  ca-   atramformerfeedingadiodefexperienceilldicatestbat  this  ratio  should 
pacity  between  windings) .    ^  2.7-3.0.    When  the  transformer  is  used  in  the  plate  of  a  modulator 

8.  To  prevent  detuning    tube,  the  ratio  should  be  2.5-2.6. 
with    avc,    those    tubes 

which  have  avc  may  be  supplied  with  an  unbypassed  cathode  resistor  of  about  100  ohms. 

9.  Tuning  usually  is  accomplished  by  means  of  iron  slug  to  increase  the  stability. 
A  typical  design  is  given  in  Table  4  and  Fig.  5. 

Each  coil  adjustable 
from  4.2  to  S.4pl)5s 
Nf=:lO.7  me 


AM 

FM 

Frequency 

0  455  Me 

10.7  Me 

L!  =  Lz 
adjusted  by  iron  core  from  

862  pA 

4.2  tih 

to 

1395  jih 

«.4  jih 

Q     ..                      .                    ... 

68-77 

88 

K 

0.70  approx. 

0.80 

Kc           '    *                              
Gain  

43  db 

28  db 

We                                           ..... 

13.5  kc 

215  kc 

W+Q     .                            .      .                            .... 

36  kc 

540  kc 

WM 

2,66 

2.50 

We 

+250  v 
FIG.  5.     Typical  0.455  and  10.7  Me  I-f  Stage  for  Combined  A-m  and  F-m  Receive 


7-42 


TACUUH^TUBE  CIKCUIT  ELEMENTS 


TOEF0L  REUITIOHS  IOR  HIGH-(?  CIRCUITS. 
Selectivity  oya  be  expressed  in  two  ways: 
\#;  With  ecnstant  input  aa 

^  Output  Toitaga  at  any  frequency  /  (or  bandwidth  /»  —  2}/  —  /ol) 
"*  Output  voltage  at  the  resonant  frequency  /Q 

«*  &  number  less  than  unity 
(I)  Wish  ean0Uiftt  output  as 


. 

*"  5  *" 


vohagg  at  any  frequency'  /  (or  bandwidth  /»  ~  2|/  — 

Input  voltage  at  the  resonant  frequency  /o 
-=  a  number  larger  than  unity 


(1) 


(2) 


resonant  frequef»cy 
band  wfdJh  at  any  value  oCA 
(same  units  asjg) 
h  at  a  <&'s 


K*W/t 


ivcl 

^rz  |     **T* 

o[  '« 
,  <at    c 

'   i-O        t 

<t»,    |8 

—  >w— 

> 

>t,        «* 

» 

!         f              A/    ^ror5M»w:"tafraal» 

%     I*     =7^-  ffocdemblerBnedoirctftts 
T          XsjJ          »adK=lCe 

wkSth  (j^/2)  multiplied  by  Qj6^ 
1012            2           8 

1 

Vali 

»s  sf  haH  bar,d 
3         "  2 

„ 

A*£* 

//• 

<^~~ 

"*N- 

1    * 

I     - 

\, 

ir 

1  5: 

\ 

-2 

101 

-€ 

( 

15- 

-4 
^ 

-6 
-7 
-8 

-9 

\ 

7 

A 

\ 

^^vity,  A,  d  m«^e  do«fc4e-tuned  stage  i& 


R^onaiat  Ireqoeacy 


to  8%2*1  3  db  greater  than  resonance       A/ 
3  db  greater  than  the  signal  applied  at 


(3) 
>  resonant 


WIDE-BAND  I-F  AMPLIFIERS  7-63 

2.  Selectivity  -  S  =  Signal  at  /  (or /-/,)  =  i_ 
Signal  at  /0  A 

(See  Section  6,  article  1,  for  universal     (5cx) 


h 

3.  Gain  per  Stage 

G  =  gmQX  (6) 

4.  Stage  Gain  Multiplied  by  Bandwidth  at  3  db 
(a)  One  stage 

G  Au  =  £m/C  (7) 

(6)  n  stages 


»  =  — —  —  (approx.)  (8) 

Vn     c 
B.  Double^tuned  Stage  (Fig.  6). 

The  following  formulas  hold  when  LiCi  =  L^C^  Qi  —  Qzi  £  —  coefficient  of  coupling; 
Kc  =  critical  coupHng  coefficient. 

1.  When  K  ^  JTC,  then,  for  each  stage, 

.  Resonant  frequency _^1  >  n 

Bandwidth  to  signal  1  db  greater  than  resonance      /c^  ~ 
Bandwidth  to  signal  20  db  above  resonance  _  WSQ      ^ 
Bandwidth  to  signal  6  db  above  resonance        Ws  ~~  ^ 

2.  Selectivity,  S 

(a)  K^Ke 

_  Signal  at  /  (or  /  -  /o)  ^  j_ 
Signal  at  /o  A 

(11) 


(See  Section  6,  article  5.) 
(5)  K  =  K 

S  = 


3.  Gain  per  ^toge,  (? 
(a) 


(6)  js:  =  xc 

G  =   V^mQX  (U) 

4.  5to^€  (?azn  Multiplied  by  Overall  Bandwidth 
(a)  One  stage 

G  AOJ  *  0.707  —•    where  C  is  the  actual  timing  C  and  is  Vs  of  total  C     (15) 

C          (approx.) 
(&)  n  stages 


13.  WTDE-BAKD  I-F  AMPLIFIEI^ 

The  amplification  of  video  signals  such  as  are  used  in  television  and  pulse  communica- 
tions requires  band  widths  wider  than  200  kc.  Hence  techniques  have  been  developed  to 
design  much  wider  band  amplifiers  of  high  overall  gain.  One  special  problem  arising  is 


VACtJUM-TUBE  CIRCUIT  ELEMENTS 

tUft  much  of  their  use  is  for  dually  displayed  information  where  the  phase  distortion 
intiwfemd  by  a  rectangular-topped  selectivity  curve  cannot  be  tolerated.  For  this  rea- 
mm,  the  *dam  of  the  selectivity  curve  must  be  somewhat  rounded.  ^ 

&LTBKJI  A1WE  DESIGNS.  Wide-band  amplifiers  may  be  designed  using  (1)  syncnro- 
aatttfr  ti*B©d  Miicfe-umed  ctrcuite,  (2)  double-tuned  circuits  (loaded  either  on  one  or  both 
•idea.  (3)  9*agger~tui»d  amplifiers,  or  (4)  inverse-feedback  amplifiers.  Stagger-tuned 
amptiftm  eonwrt  of  a  sm^-tiuwi  circuits  of  poor  skirt  selectivity  which  are  tuned  to 
diffmnt  f  reqwums  to  that  the  pe&k  of  oee  circuit  tends  to  fill  in  a  deficiency  of  the  others. 
F«r  touaee  a  ptacger-ttiaed  pair  consists  of  two  single-tuned  stages,  one  of  which  is 
pmk#d  at  a  frequency  higher  than,  the  other  lower  than,  the  center  frequency  of  the 
•mpli&r.  la  in^m-feedback  amplifiers  a  fraction  of  the  output  of  a  single-tuned  syn- 
oiarmmm  amplifier  is  fed  back  in  degenerative  phase  to  the  input.  More  voltage,  there- 
fore, ia  fed  Imrk  at  rfjson&nc*?  tlmn  o&-re80nance.  Consequently,  the  overall  gain  is  reduced 
mew  at  r®mm®&@®  than  off,  and  the  nose  of  the  selectivity  curve  is  rounded. 

FIGURE  OF  MK11T,    The  Sgure  of  merit  of  a  wide-band  amplifier  is  defined  as 

Stage  gain  times  overall  bandwidth  —  G  A&? 

w^ft  the  band  width  Ju^  means  the  band  width  for  gains  3  db  below  the  peak.  The  3-db 
land  width  is  ehoeen  because  it  makes  the  mathematics  easier,  it  approximates  the  noise 
hfttt'iwidth  erf  the  rweiwr,  and  so  facilitates  signal-noise  comparisons.  Furthermore, 
with  the  usual  ©owpting  circuits,  the  rise  time  of  pulses  is  quite  simply  connected  with  it. 
Defhunc  m*  tiiM  m  th*  time  required  for  the  response  to  the  step  function  to  increase 
frori  10  to  90  per  cent  of  its  final  value,  then 

0.7 
Hise  time  =*  — 

Tlmm  »  10-Me  (S  db  dbwa)  i-f  &mpliSer  would  have  a  minimum  pulse  rise  time  of  0.07  jus. 
(For  *  €M»aplH@  eritemts  of  a  puke  amplifier,  percentage  of  pulse  overshoot  must  also  be 
<tMQtaidtf*d. ) 

For  m  ^mgl^^tage  single-tuned  circuit  the  gain  of  an  amplifier  is  (Z  is  the  plate  load 

^m    ^  gm 

—  — -  — 


(17) 

i  is  aho  eq.  (7).  Tb*  fawelor  C  represents  the  total  capacitance  in  the  plate  circuit, 
infiudi&g  tube  output  and  input  capacitances  and  the  capacitance  of  circuit  components 
to  fproufid. 

Since  double-tuned  nroaits  divide  the  total  tube  and  circuit  capacitance  between  two 
tuaed  (ircuita  they  use  fe«s  capacitance  per  tuoed  circuit  and  have  a  higher  figure  of  merit. 
For  equal  Q*»  this  m  VI  g»/C,  asd  for  ®m  side  loaded  it  is  2gm/C. 

STWCHROKOIFS    STH GLE-TTJHED    CIRCUITS.*     The  sim- 
OveriS      ptkity  a®d  stafeiity  of  an  amplifier  ma<fe  up  of  single-fcuned  circuits 
@|         aU  tuned  to  resonance  oommeiid  it  to  the  designer,  and  most  2-  to 

j  si y       S-McHwkle  amplifiers  are  of  this  type.    For  really  wide-band  high- 

i  Amplifiers  pan  mmpMfkra  the  decrease  in  bandwidth  arising  from  multipli- 
cation ol  the  successive  selectivity  curves  makes  this  type  unsuitable. 
Table  §  shows  th«  overall  bandwidth  of  it-stage  amplifiers  in  terms 
ol  the  %&m®e  bandwidth.  It  siiows  that  a  nine-stage  synchronous 
^^jie-t^acd  amplifier  with  an  overall  bandwidth  of  4  Me  requires 
a  singliHstAge  baadwMth  of  14.3  Me  with  its  correspondingly  low 
g»wa.  It  eaa  be  shown  that  the  maximum  overall  bandwidth  of  syn- 
Aroamii  ^agle-tuaed  circuits  occurs  for  a  mean  stage  gain  of 

g  -  V*~=  4.34  db 


DOOTI^-TUHED  CIRCUITS.     Double-tuned  circuits  have  a 
ita««  in  the  figure  of  merit.     However,  the  large 
-^^jfeaents  (three  per  stage)  and  the  criticalness  of 
i)  make  desagn  and  maintenance  most  difl&cult.    For  this 
fern Jaeea  wxsd  vwry  little. 

CmcuiTS.    WaBman  defines  an  exact  stag- 
geoiaetricaBy  centered  at/*  as  consisting  of  i 
*"  J  ~  l/Q  and  :'  *         -        - 


WIDE-BAND   I-F  AMPLIFIEBS 


7-65 


(for  one  triple)  (6)  one  single-tuned  circuit  of  dissipation  factor  5  —  A///  centered  at  /Q. 
d  and  a  are  plotted  as  functions  of  5  =  A///  in  Figs.  7  and  8.  (Note:  The  values  of  d  and 
a  for  a  pair  are  not  the  same  as  their  value  for  a  triple.) 

The  stage  gain  times  overall  bandwidth  of  (a)  a  single-tuned  stage,  (b)  an  exact  stag- 
gered pair,  and  (c)  an  exact  staggered  triple  is 

G  S  =  gm/C 

However,  the  same  3-db  bandwidth  AOJ  is  obtained  in  the  first  case  (a)  for  one  tuned 
circuit;  in  the  second  case  (b)  for  two  tuned  circuits,  and  (c)  for  three  tuned  circuits. 


/• 

0   fi 

j 

/ 

1  9 

/ 

i  R 

/ 

/ 

/ 

X 

1  6 

/ 

^^"^ 

^ 

1  ^ 

/ 

S 



^ 

^t*** 

1.4 
1.3 
1.2 
1.1 
1.0 
0.9 
0.8 
0.7 
06 
0.5 
0.4 
0.3 
0.2 
0.1 

Ex 

act  value  o 

| 

ee-. 
<S 

?/ 

^ 

^ 

1 

J-d 

^ 

^^ 

1-h. 
Asym 

353  1 
ptotic 
fors 
sof^ 

' 
valu 
mall 

/ 

/ 

s". 

^ 

of  a. 

jfaluq 

/ 

^ 

^ 

2^ 

s 

/ 

^ 

^ 

„*•-'"* 

/ 

s 

.  

^ 

/ 

^< 

•^ 

/ 

/ 

^^ 

^ 

0.7075 
Asymptotic 

valu 

/I 

^ 

Exi 

ct  va 

ue  of 

J 

of  d  for  small 
values  of  S-\_ 

s 

^ 

^ 

st 

V 

S 

/ 

s 

/ 

S 

/ 

/ 

f 

0    0.1  0.2  0.3  0.4  0.5  0.6  0.7  0.8  0.9   1.0   1.1  1.2   1.3  1.4  1.5  1.6  1.7   1.8   1.9  2.0 


PIG.  7.     Design  Curves  for  an  Exact  Flat  Staggered  Pair 

The  algebra  in  the  analysis  of  these  circuits  is  quite  involved,  but  Wallman  has  reduced 
the  design  to  charts.  Given  the  tube  type  and  the  general  layout,  the  gm  is  known  and 
an  estimate  of  C  is  made.  This  determines  the  product  of  swgknBtage  gain  and  ov&raJl 
bandwidth,  i.e.,  G  Aw  =  gm/C  of  the  pair  or  triple. 

The  overall  bandwidth  of  the  amplifier  can  be  obtained  from  the  following  approximate 
relations  _ 

_,.,,,          .  Bandwidth  of  one  pair  (i.e.,  Aw) 

1.  Bandwidth  of  n  pairs  =  -  4-^  - 

1.1      n 


2. 


Bandwidth  of  n  triples  : 


Bandwidth  of  one  triple  (i.e.,  Ao?) 


1.06 


7-66 


VACUUM-TUBE  CIRCUIT  ELEMENTS 


23 
2LJL 

2.1 

13 

U 
UB 

1.4 

3-3 
UZ 

1*0 

ex* 
CUB' 

OJ 

0,6 

0J2 

ft  1 

1 

/ 

x 

x 

x 

X 

^ 

X 

^ 

^ 

Ei» 

:tv*l 

je  Of 

a-^ 

x 

^x 

^ 

^ 

X 

.<^ 

X" 

^ 

x 

^^ 

^ 

Asymptotic  v 

)ue 

j 

S 

^ 

X"" 

of  Of  for  sma 
values  of  5 

^ 

\J^ 

+> 

^ 

r"! 

^ 

^ 

^ 

^ 

^ 

^ 

^~~~ 

faym 
tfdi 
rtvu 

^Dtk 

of  srr 

wk» 

*9 

^ 

^ 

^^^, 

^^ 

*~~~ 

.  —  • 

4 
1 

V 

^* 

^  — 

\       \ 

~Lx 

x^ 

^< 

-Exj 

ct  va 

ueof 

d 

\ 

<^ 

ss^t 

•^* 

A 

X 

x 

^ 

<^ 

\ 

, 

0    CU  02  08  CL4  05  06  0.7  0.8  03  1.0  1.1  1.2  13  IA  1.5  1.6  1.7   1.8  1  9  2 

Design  Cianros  for  &a  Exact  Staggered  Triple 


Cwwtrvtfcr,  tfe«  oreral!  fe^kdwidtfe  and  oT^all  gain  being  known,  the  number  of  stages  as 
vsaiA  p«siioriaifcaee  can  be  determined. 


Tte  •t«ci 


1(^  db  '^^^  <»  30  Me  is  wanted.    6AC7's 
total  capaciUnw  ^  25.5  ftpl  aje  to  be  used. 


»  108  t  -  12  «fb,  ysat  ie,  a  voltage  »aia  <^  4  times. 

-  X  1013  -  3^  X  10» 


^^Mii  tiivii  *re  ftmsd  ta^  H«.  Sw 


WIDE-BAND   I-F  AMPLIFIERS 


7-67 


II 
s  § 

li 


.£ 
"? 


7-48 


VACUUM-TUBE  CIRCUIT  ELEMENTS 


Use  Hftatiaer,  therefore,  •  made  tip  of  three  triples,  each  of  which  consists  of: 

1.  One  tirewt  tuned  to  30  X  1.21  -  36.3  Me  and  having  a  Q  of  4.65. 

2.  Otw  eireuit  tuned  to  30/1.21  -  24.8  Me  and  having  a  Q  of  4.65. 

3.  €fm  ctreuil  tuned  to  30  Me  ami  having  a  Q  of  1/0.443  =»  2.26. 

»  Q  «  «Cft  (K  in  thw  ewe  k  the  equivakaat  resistance  across  the  tuned  circuit),  then 
Q  4.65  __  ^  79Qw  for  the  36.3.MC  circuit 


*  2*  X  24.S  X  10*  X  25,$  X  I® 


—   «  1150w  for  the  24.8-Mc  circuit 

i— 12 


2.2® 


X  30  X 


X  lO 


___  M  470<d  for  the  30-Mc  circuit 

i-19 


Tbt  awis^w  »  sfcowa  in  Fig.  9.    Actuallj*  the  physical  vmloes  of  the  shunting  resistors  are  higher 
this*  a»le«kf«d  bwmw*-  ^  the  effort  of  tube  loading  and  of  the  finite  Q  of  the  coils  which  effectively 
thuat  tfc*  j*>nwa  iwwtor  to  the  eakuUted  value.    The  final  value  of  these  resistors  must  be  deter- 
" 

ttac*  Q*  ^^  triple  is  determined  from  the  relation 


tk*t  tbe  rr«^»  fo 


i  triple  can  be  tibulited  as  follows: 


A.M« 

0 

A/,  Me 

Shunt 
Resistance,  ohms 

O)  54.3 
f2)  24,8 
«3)  M,© 

4.65 

4.65 
2.26 

7.9 
5.4 
13.3 

790 
1150 
470 

tables  which  extend  tlie  dmgn  to  quadruples,  and  so  forth,  called 

IHTERSE  FEEDBACK  I-F  AMPLIFIER  CIRCUITS.  Typical  feedback 
drain*  are  abown  isa  Fig^  ICte,  1(M>,  ami  lOc,  These  are  drawn  without  regard  to  d-c  po- 
MkBtial^  TfciM  ttoe  grkli  imiai  foe  protected  from  the  plate  voltages  by  blocking  condensers 
or,  pffdffmNy,  IraBtforswrs  hmvisg  uaity  coupling. 

Tbe  f«4b*ck  Aaiia^  of  Fig,  ICki  is  ol  little  use,  despite  its  high  gain  times  bandwidth 
factor,  foeewist  ila  gpm  caaaot  be  ecmtrolled.    Figures  106  aad  lOc  show  respectively  a 


fVWv 


la  each  case  the  voltage  which  controls  the 


pair  ami  ft^dbaefc  tri^e. 

ii  «$|)1M<1  to  the  ir^  tub©. 

A  ««»^»le  i-f  asapiii&r  mm®  0AK5's  mad  imv«nse  feedback  is  shown  in  Fig  11 
TW»  MqpMw  ©mwts  €^  *w^  triplati  mad®  up  re^ectiTely  of  tubes  V-301  V-^02  V-303 
d  V^W.  V-^S,  V-306,    Gaia  octroi  is  appfei  to  V-301  and  V-304.    This  alnplifier 
60  Me  n^i  has  aa  overall  gmia  of  a^jroximately  90  db  arid  a  bandwidth  at 

10  Me. 

«f  featecfc  amplifcrs  U  not  m  readUy  comfmted  as  that  of  stagger-tuned 
Hwwifof®  M  Im^d  to  a  greater  extent  on  test  and  experience 
F«r  am  ««^^  ««Mdy  of  f«*sdb»ek  ampiife^  ^  rel^enoe  9,  10,  11,  13,  and  14. 


WIDE-BAND  I-F  AMPLIFIERS 


7-69 


f-70  VACCUM-TUBB  CIRCUIT  ELEEMNTS 

*    STAGGSR-TtJimD  AMPLIFIERS  VS.  INVERSE  FEEDBACK  AMPLIFIERS.     The 

etew  between  a  stagger-tuned  aad  an  inverse-feedback  amplifier  depends  largely  on  the 
omim*t*xa<!*is  suritmsadiiig  it«  use-  There  is  no  fixed  answer  as  to  which  is  best  for  all 
condition*.  The  difference®  are  outlined  below. 

Fifmr«  ©f  Mftiit.  Both  th«e®  types  ol  amplifiers  have  been  shown  to  have  a  much 
higher  6gura  of  merit  than  the  synchronous  single-tuned  circuit  type.  Theoretically  the 
tov*r«e»f««db*c!t  amplifier  can  have  a  higher  one  than  the  stagger-tuned,  but  it  is  difficult 
to  actssrrf  tb»  in  prirtiee,  »  thmt  they  may  be  assumed  equal. 

G*is  C«atr«*L  Gain  adjustment  may  be  desired  (1)  to  change  the  output,  (2)  to  change 
UM  gain  with  time  (pan  time  control,  gtc  in  radar  applications),  (3)  to  reduce  interfer- 
«o«*.  (4)  for  automatic  gain  stabilisation  (ags),  (5)  for  automatic  gain  control  (avc). 

A  ttaiHpv^UuMd  amplMiar''®  bandwidth  being  practically  independent  of  the  tube's 
I «,,  gain  control  can  be  applif-d  to  ^115'  tube. 

AM  tavern-feedback  amplifier  emu  use  for  gain  control  only  those  tubes  which  are  not 
in  tlw  feedb^^k  coam.  Otherwi^,  the  bandwidth  will  change  with  gain. 

Gaia  Y*rimti®a  witli  g w.  The  gain  varies  more  slowly  with  gm  in  a  feedback  amplifier 
(approximately  as  i»^!  than  in  others  (linearly  with  gm).  This  means  that  the  gain  of 
msBpiiSwi  not  equipped  with  automatic  gaia  stabilization  (ags)  will  be  less  susceptible  to 
voltage  rhaBgf,  tub®  ageing,  and  tube  replacement  if  of  the  feedback  type. 

Kffact  ^  Rc^ac«m3emt  Tufees  on  Bandwidth.  In  a  feedback  amplifier,  the  resonant 
frequency*  and  thfrefore  th«  overall  bandwidth,  is  less  affected  by  replacement  tubes  of 
differing  cmpadtsucje,  Howmsr,  the  bauadwMth  is  more  affected  by  tubes  of  differing  gm. 
;  Frequeacj.  An  iDveree-feedback  amplifier  has  all  its  circuits  tuned  to  the 
of  the  pass  band. 

Tuaiag.     In  a  stag^r-tuaed  amplifier  the  tuning  of  one  stage  is  in- 
'  the  taking  01  the  other  stage. 

BIBUOGRAPBnr 

1,  Dfemrn,  C.  E%»«a^  WMtfe  fWtora  f®r  Cswakde  Tuned  Circuits,  Electronics,  July  1941. 

2,  Sf*a3sl'Ai»>c»   W,  E»#  Jr.,  DS^EB  <3l  Supcrfeeterodya®  Intarniedi&te-frequency  Circxiits,  RCA  Rev.. 

Apr*]  1WK 
S»  FnwaeMMk,  E.  1*.,  IT*B  ol  F«e«fiM^!k  lo  Compensate  fca-  Vacuum  Tube  Input  Capacitance  Variation 

with  Bi»,  /We.  /»JLJL|  NGrwrnbes-  1 938. 
4.  Wl««ei*r,  H*  A,,  aand  J.  1C.  Joteim,  High  fidelity  Receivers  with  Expanding  Selectors,  Proc* 

IT.  H.  A.«  Wide  Bfetml  Amplifeefm,  jmp«r  presented  at  joint  meeting  of  LR.E.  and  A.LE.E. 


m  New  York,  Apri  IS,          . 
ft,  Ffirt«r,  R  £.,  »ad  4F,  A,  Eiyalcsm,  lat^Hsediatc-lrwiiiaaey  Values  for  Frequency-modulated  Wave 

Rmvivft,  Pr^c,  /JU^,  Octdfcw  1941. 

7.  Zwwrto,  V.  K.f  «ml  G.  A.  Mwton.  TfimnMi,  Ch»pt«r  17.  John  Wiky  (1&40). 
*•  S^^Sf11*  HmrirvS«M»8r-ti»«4  I-f  AaBi*S«ra»  R*sii»tiosi  laboratory  R^ort  524,  NDRC  Div.  14. 
•,  TJM  T^WFJ  of  M  A»|^&»«  with  Ke«j*iive  Feedback,  Canadian  N&tiooal  Research  Council 


. 

St.  UwwW^E.  N^  felwM^a  o«  BreawS-feaad  Feedback  I-l  Ampfifiers*  CRG  Report  93,  Naval 
ri  Latbcrstory,  Oct.  22,  1945. 


,        .      ,          . 

IL  ^^W™*  W"  E^a^9^  fm  ^v®^*^  AmpBfiats,  CRG  Technical  Memorandum  217,  Nov.  26, 
W**ie  Barf  Am^^Nwt  for  T«j®rfeaw, 
TAJtar  attd  Waflwam,  FarwiMi  Tulw  4 

f»,   JM'CCjrtWW-KliS, 


IZ  W**ie  Barf  Am^^Nwt  for  T«j®rfeaw,  J^V^tr.  I.&.E.,  July  1939. 

ii.  TAJtar  attd  Waflwam,  FarwiMi  Tulw  4»»iaftM^,  Had.  L*bo«itot7  Series  No.  18,  Chapters  4,  5,  and 

f»  'Cjr- 


MODULATORS 

ByJ.  E,  Yimce 


tto     WMISS  wiw^  tfee  amplitude,  <»•  other  eiiaract^Ktic,  <^  a  wave  is 
wcm®  rahie  of  anotl^r  wave.    The  firsfe  wa^e,  which  is 
llie  "earner  wave";  the  second  is  called  the 


cftrrier  fr^mttmcj  m  th*^  lrmm«ii<iT  c^  the  carrier  wave. 

i«»vtiHifrft«nMMgr  tmn«is  o®  ea^fet  side  <^  the  earner  frequency  within  which 

of  modoli^m.    In  amplitude 


.  . 

^t^wytfe  o|  a  trmmmiinsd  ^delmiMi  is  anally  no  greater  than  the  band- 

***    «  a  f  ^^-mcHJulated  sgnal  is  de- 

^ 


taHMri  Igr  bodi  tbi 


TYPES  OF  MODULATION  7-71 

Percentage  modulation  (amplitude  modulation)  is  the  ratio  of  the  difference  between 
the  maximum  current  (Zm)  when  a  signal  is  impressed  and  the  maximum  unmodulated 
current  (I«)  to  this  maximum  unmodulated  current,  expressed  in  percentage,  or  M  = 
100(7 'm  —  It} /Is-  For  less  than  100  per  cent  modulation  this  may  optionally  be  expressed 
as  the  ratio  of  half  the  difference  between  the  maximum  and  rninimurn  amplitudes  of  a 
modulated  wave  to  the  average  amplitude,  expressed  in  percentage. 

Percentage  modulation  (frequency  modulation)  is  the  ratio  of  the  frequency  swing  when 
a  signal  is  impressed,  to  an  arbitrary  frequency  swing  which  is  defined  as  100  per  cent 
modulation,  expressed  in  percentage.  The  frequency  swing  defined  as  100  per  cent  modu- 
lation is  different  for  various  services.  For  f-m  broadcasting,  for  example,  75  kc  is  used. 

14.  TYPES  OF  MODULATION 

A  single-frequency  current  wave  can  be  expressed  as 

i  =  A  sin  («i  -f  0)  (1) 

where  A  is  the  amplitude,  co/27r  the  frequency  (when  constant),  and  &  the  relative  phase. 
If  any  of  the  three  independent  magnitudes,  A,  co,  or  0,  is  slowly  varied  (slow  in  com- 
parison to  co/2x)  the  wave  is  said  to  be  modulated.  The  three  cases  are  called  ampli- 
tude, frequency,  and  phase  modulation,  respectively. 

If  w  and  0  are  held  constant  but  A  is  varied  sinusoidally,  so  that  A  =  Jt0(l  +  m  sin  t^t) , 
eq.  (1)  becomes 

i  =  jto(l  4-  m  sin  «ii)  sin  (^  -f  #)  (2) 

which,  when  expanded  trigonometrically,  gives 

i  =  A0  sin  (tat  +  0)  -f  ^  {sin  [(w  +  «n)t  -f  &}  +  sin  [(«  -  «*)*  +  03}  (3) 

The  quantities  in  the  bracket  of  the  second  term  on  the  right  represent  sum  and  difference 
frequencies  of  the  two  original  frequencies.  They  are  the  sidebands.  Comparison  of 
these  two  equations  shows  that  a  wave  of  a  single  frequency  and  periodically  varying 
amplitude  is  mathematically  equivalent  to  a  wave  of  constant  amplitude  and  frequency 
and  a  pair  of  sidebands. 

Roder  has  shown  (Proc.  I.R.E.*  Vol.  19,  2145  [1931])  that,  if  A  and  0  are  constant  but 
w  ~  o>o(l  +  kf  cos  jo£) ,  the  current  is  (let  c«>o£  =  tat  +  6) 

i  =  AQ  sin  woi  +  Ji(m/)[sin  (coo  -f  M)*  —  sin  (too  —  fi)t] 

—  JjOw/Hsin.  (o>o  -f  2^)2  —  sin  (coo  —  2;*)fj 

-f  /3(m/)[sin  (wo  4-  SM)*  -  sin  (wo  -  3jn)«     (4) 

where  m/  —  fe/co/ju  so  that  m/  is  the  ratio  between  the  maximum  frequency  shift  and  the 
audio  frequency;  also  Jn(ri)  means  the  Bessel  function  (see  Section  1,  article  14)  of  the 
first  kind  and  nth  order  for  the  argument  m.  In  this  expression  there  are  theoretically  an 
infinite  number  of  sidebands,  although  the  amplitude  of  all  those  of  higher  order  than  the 
first  is  usually  negligible. 

If  A  and  o>  are  constant  but  6  —  #0(1  +  KP  sin  p£)f  Roder  gives  the  current  as  identical 
in  form  with  eq.  (4)  except  that  m  =  KP6Q.  Here,  again,  there  are  theoretically  an  infinite 
number  of  sidebands,  but  only  those  of  the  first  order  are  of  importance  or  of  appreciable 
amplitude. 

Except  for  some  very  early  and  ineffective  attempts  to  use  frequency  or  phase  modula- 
tion, amplitude  modulation  was  used  exclusively  for  communication  and  broadcasting  up 
to  1936.  Exploration  into  the  higher  radio  frequencies  and  the  adaptation  of  these  fre- 
quencies to  wide  use  for  numerous  services  has  brought  frequency-  and  phase-modulation 
techniques  into  great  importance,  since  the  wider  frequency  bands  required,  if  they  are  to 
be  used  effectively,  become  available  as  the  usable  frequency  spectarum  is  extended  into 
the  hundreds  and  thousands  of  megacycles. 

AMPLITUDE  MODtfLATION.  Since  the  e2  term  of  the  power  series  of  development 
for  current  in  a  non-linear  circuit  gives  rise  to  the  term  sin  c^t  sin  w£,  characteristic  of 
modulation,  it  follows  that  modulation  occurs  whenever  any  of  tfie  circuit  parameters  c»ar# 
with  instantaneous  voltage  (or  current).  In  particular,  if  it  is  assumed  that  a  single  high- 
frequency  voltage  epf  —  E*  cos  (&JL  +  0»)  is  introduced  in  the  plate  circuit  of  a  vacuum 
tube  and  a  varying  audio  voltage  eg  —  2  Esk  cos  (caj^  4-  6k)  is  impressed  on  the  grid,  then, 

I 


7-72  VACUUM-TUBE  CffiCUIT  ELEMENTS 

fa*M  Be«*k»  ft.  mkie  21,  «q.  (17),  the  tarn*  having  frequencies  near  that  of  aJZ*  are 

»V»  -  : 


*»* 


I  - 
L 


COS  (ttj$  — 


J 


bt  wiittaA  (WHOM  t&* 


as  iU^tmted  there  IE  eq.  (22).    This  may  optionally 
<rf  *<,4jk,  and  *(._«  negHgible  as  is  usual  in  practical 


tfct 


Th® 


(7) 


ia  the  brmrket  ia  aimply  the  maximum  value  of  the  audiofrequency  plate 
ar«  ahown  in  Fig.  1. 


A.-  Signal  Curraisl 


One  Component  of 

the  Second  TerrQ 

c^  Equation  (5) 


Tolal  of  Equation  (5) 

for  Two  Sine  Wave. 

Voltages 


The  Secood  Term 
of  Equation  (5) 


Equation  (5)vsfjlh 
One  Part  of  the 
Second  Term 

-    -    -    -  Eliminated 

F.-  Carrier  *mf  Om  S^k  Band 

F»fc  I.    Cemfeiaa^®  ef  Sine  Wave  Carrier  aad  ^gaal  Curr^iis 

METHODS  Of  PKOI>UCIIf G  AliHJTirBl  MODULATIOK.    Systems  of  modula- 

"*--•»"•••••    s  imto  turn  daa®^: 

kti  t^  iai|j«4aiice  of  *  r-f  oeallator,  amplifier,  or  combination  of 

.  l§  twwr*iNsl  tsgf  thfe  tao^ilatiiig  wave. 

_t.  8v*t«m!»  feaa^ym^j  a  eo^taBt-impedaBce  r-f  oscillator,  or  amplifier,  having  a  var- 
*a  aeriea  w^Jbt  it  in  whidh  the  variation  o!  the  magnitude  of  the  series 


D*-  Carrier  wtd  Bi^fe  Skie  Bamh 


GRID  MODULATION  7-73 

impedance  is  controlled  by  the  modulating  wave,  thereby  controlling  the  input  to  the  r-f 
unit. 

The  first  method  of  modulation  may  be  accomplished  by  varying  the  grid  bias  of  the 
r-f  oscillator  or  amplifier  if  a  triode  is  used  or,  alternatively,  varying  the  suppressor  or 
screen-grid  potential  if  multielectrode  tubes  are  used. 

The  second  method  depends  for  its  operation  on  the  introduction  of  the  modulating 
emf  in  series  with  the  plate  power  supply  of  the  modulated  oscillator  or  amplifier.  Since 
appreciable  amounts  of  power  are  required  to  accomplish  modulation  in  this  manner, 
auxiliary  tubes  are  employed  as  modulators. 

15.  GRID  MODULATION 

The  term  grid  modulation  is  commonly  used  to  describe  modulation  systems  of  the  first 
class  mentioned  above.  Modulation  is  effected  by  varying  the  d-c  grid  bias  voltage  or 
suppressor  grid  voltage  at  an  a-f  rate,  by  connecting  the  modulating  source,  such  as  the 
output  transformer  of  an  audio-amplifier  stage,  between  the  bias  source  and  the  r-f  stage. 
Since  the  grid  input  resistance  is  high,  the  audio  amplifier  need  be  capable  of  only  a  few 
watts  output.  The  power  required  is  still  further  reduced  in  some  designs  by  the  use  of  a 
tube  in  the  r-f  circuit  which  is  capable  of  large  plate  currents  without  exceeding  zero  grid 
bias.  In  this  case  the  a-f  amplifier  looks  into  a  circuit  of  substantially  infinite  impedance. 
The  operating  bias  for  the  r-f  stage  is  determined  as  follows:  The  r-f  tube  is  first  biased 
to  cutoff,  with  no  r-f  grid  voltage.  The  r-f  grid  voltage  is  then  increased  until  the  knee  of 
the  saturation  curve  is  reached.  Then,  with  a  constant  r-f  grid  voltage,  the  d-c  grid 
voltage  is  increased  until  the  plate  current  is  substantially  zero.  If  ei  is  the  d-c  grid  volt- 
age for  maximum  output,  and  ez  is  the  d-c  grid  voltage  for  zero  output,  the  operating  grid 
voltage  should  be  (eg  +  ei)/2.  The  peak  a-c  voltage  required  from  the  audio  stage  to 
produce  complete  modulation  is  (e%  —  ei)/2.  Because  of  the  relatively  low  efficiency 
obtained,  grid  modulation  methods  are  generally  applied  only  where  the  amount  of  mod- 
ulated power  required  is  very  small.  Higher  powers  are  sometimes  obtained  by  following 
the  grid  modulated  stage  by  r-f  amplifiers.  The  outstanding  advantages  of  grid  modula- 
tion are  its  simplicity  and  cheapness,  which  make  it  attractive  for  small,  light-weight  appli- 
cations, such  as  portable  or  airborne. 

POWER  AND  EFFICIENCY  OF  GRID -BIAS-MODULATED  AMPLIFIERS.  The 
following  discussion  may  be  applied  in  determining  peak  power  requirements,  efficiency, 
driving  power,  etc.,  of  class  B  r-f  amplifiers  as  well  as  bias-modulated  amplifiers,  since 
the  tubes  operate  under  the  same  conditions  in  each  ease.  The  peak  power  output  capa- 
bility of  the  tube  must  be  four  times  the  carrier  power  if  complete  modulation  is  to  be 
obtained.  This  follows  from  the  fact  that  the  output  current  and  voltage  must  be  doubled 
on  peaks  of  100  per  cent  modulation.  The  efficiency  is  proportional  to  the  percentage  of 
modulation,  and  it  is  of  the  order  of  20  to  33  per  cent  for  normal  carrier  or  unmodulated 
conditions.  The  ratio  of  driving  power  of  the  r-f  exciter  to  carrier  power  of  the  modulated 
stage  should  not  exceed  1  :  10,  since  the  exciter  must  maintain  a  constant  r-f  voltage  on 
the  grid  of  the  modulated  tube.  The  following  formulas  show  the  current,  voltage,  power 
relations,  and  effect  of  the  variation  of  the  percentage  of  modulation. 

Let  Wp  —  peak  instantaneous  power  output;  We  =  carrier  output  power;  WA.  =  aver- 
age power  output,  100  per  cent  modulation  (used  in  determining  heating  effect) ;  ipc  =  peak 
plate  current  averaged  over  an  r-f  cycle,  100  per  cent  modulation;  ipm  =  average  plate 
current,  unmodulated;  fp  =  maximum  instantaneous  plate  efficiency  averaged  over  an 
r-f  cycle,  100  per  cent  modulation;  /c  —  average  plate  efficiency,  carrier  unmodulated; 
F  =  modulation  factor  =  M/IQQ;  WLP  =  plate  loss  averaged  over  an  audio  cycle,  100 
per  cent  modulation;  and  WLC  =  average  plate  loss,  carrier  unmodulated. 

Assume  that  d-c  plate  voltage  remains  constant,  and  that  modulation  is  effected  by 
variation  of  plate  current  and  efficiency.  Then: 

' 

WA  =  l-5TFc  (9) 

ipm  -  2ipc  (10) 

/,  =  fed  +  F}  (11) 

WLP  -  WLC  (12) 


7-74 


VACUUM-TUBE  CIRCUIT  ELEMENTS 


R.F. 


16.  PLATE  MODULATION 

The  Murtien  system  of  p!at«  modulation,  devised  by  Heising,  was  known  as  the  constant- 
current  avrtern.     Sw  Fig.  2.     It  derived  its  name  from  the  fact  that  variation  of  the 

power  input  to  the  r-f  stage 
is  obtained  by  using  another 
tube  or  tubes  as  an  ab- 
sorber. The  current  drawn 
from  the  supply  source  re- 
mains constant  and  shifts 
between  the  absorber  and 
useful  load  as  the  absorber 
or  modulator  impedance  is 
changed  by  the  a-f  voltage 
impressed  in  its  grid.  Be- 
cause of  the  curvature  in  the 
extremes  of  the  vacuum-tube 


Fio.  2, 


Amplitude  Modulation 


eharacwriatias  wbea  msed  as  a  modulator,  the  percentage  of  modulation,  where  the  r-f 
«Uff*  Mid  Biodiiiator  are  both  supplied  with  the  same  plate  potentials,  is  limited.  To 
obtain  compel*  modulatioa,  the  d~c  voltage  impressed  on  the  radio  stage  is  often  reduced 
by  a  swrrtw  rwafcor.  Th«  a-f  voltage  developed  by  the  modulator  is  transferred,  unatten- 
»*t«d,  t*y  n  s*iitAbb  by-pass  e&paofcor  «xmn«jeted  around  the  resistor.  It  is  usually  pos- 
\  complete  modulation  by  reducing  the  r-f  amplifier  voltage  to  60  or  70  per 

;  of  th«  modulator  voit- 

M^to^itof  To  modutated  ampfifler 

Modolatlcm  transformer 
^-*Mi —      designed  to  carry  d-c 

W  to  tfe^T  0X~  compommt  of  modulated 

I   1ft    ta^-pOWW  ^K     amplifier  plate  current 

trftmsmitlers,  -ML^ 


been  pr^atly  iiaprt>vf?d  by 


ihf  r.La^s  C 
mtsitAs  of  &  coupling 
fornser  «od  by  unnc 
of  Ki-oiiiilalor  tiibe«  iri 


following  the 

^«gt    may    be 
with  entirely  in 


Pfatt 


which    are 

limited  only  by  tl**  «sa|3^bal- 
0!  the  ¥me$nim  tubes 
in  the  roc*iul»tay  AXM! 
tfier,  witlt 

high  overall  eSctenrrj-.    Fur 

tJ^»  eaicuiaUo-n  of  feuch  eir- 

anla  the  rea*ier 

to  p.   7"-  IS. 


To  modulated  ampRfler 


fey 
the    iri<wialai<;vr 


Transformer  carries 
rnodtiiated  component 
of  modulated  amplifier 
plate  current  only 


t^rminat^d    on    the 
$id^  of  tb®  co$>- 

^  m^f  ^®'  3"    Modalfttor-amplifier  Coujiiog  Circuits 

aSas«3  »jnp}iS*r  4-c  p4*«  T^t«§e  €^Tid©d  1^-  its  plate  current.    It  should  be  noted  that, 
modulators  are  iised,  the  current  delivered  by  the  power 
la  fact,  ia  the  isa«c  of  tli©  pmshpnH  modulator,  the  current 
!wy  Iwc^*  aeeoiid  Ittnnoiue.    Special  attrition  must  be  paid 

,  «ee  Itarai^aaic  vokage  de-^eloped  aoroaa  t^  pow^r  ripply  terminals 


COMPARISON  OF  MODULATION  SYSTEMS  7-75 

by  the  modulator  current  demand  will  be  applied  to  the  modulated  amplifier  and  appear  as 
distortion  in  the  modulated  envelope.  Recommended  coupling  circuits  are  shown  in  Fig.  3. 

The  modulated  amplifier  is  operated  class  C;  however,  if  distortion  is  to  be  minimized, 
special  considerations  are  involved.  It  will  be  noted  that  the  applied  plate  supply  voltage 
to  the  class  C  stage  has  the  wave  shape  of  the  modulating  wave.  It  follows  that,  for  100 
per  cent  modulation,  the  voltage  rises  to  twice  the  carrier  level,  and  the  plate  current  must 
likewise  rise  to  twice  its  quiescent  level.  Adequate  grid  excitation  must  be  provided  to 
insure  this  increase  in  plate  current.  This  is  usually  achieved  by  driving  the  modulated 
amplifier  somewhat  harder  than  is  required  to  produce  rated  carrier  power  at  maximum 
efficiency.  In  addition,  grid  leak  bias  is  used,  and  since  the  grid  current  decreases  as  the 
plate  voltage  increases  (and  the  plate  robs  the  grid  of  some  of  its  electrons)  the  grid  bias 
falls,  the  amount  of  grid  driving  power  falls,  and,  owing  to  the  combination  of  decreased 
bias  and  driver  regulation,  the  a-c  grid  driving  voltage  increases,  thus  producing  a  condi- 
tion favorable  to  a  linear  relation  between  modulating  plate  voltage  and  r-f  output  voltage. 

The  rise  in  driving  voltage  required  to  mininiize  distortion  is,  to  some  extent,  a  function 
of  the  design  of  the  tube.  Some  types  of  tubes  require  less  increase  in  drive,  as  the  plate 
voltage  rises,  than  is  obtained  by  the  combination  of  grid  leak  bias  and  driver  regulation. 
Lowest  distortion  will  be  obtained  in  these  cases  if  a  combination  of  fixed  and  grid  leak 
bias  is  used.  The  desired  combination  may  easily  be  obtained  by  by-passing  part  of  the 
grid  leak  with  a  capacitor  large  enough  to  retain  most  of  its  charge  over  the  lowest  a-f 
cycle. 

Another  method  of  obtaining  the  necessary  variation  in  drive  is  to  modulate  the  driver 
stage.  It  is  usually  not  necessary,  for  optimum  results,  to  use  a  depth  of  modulation 
greater  than  30  or  40  per  cent  for  this  stage,  when  the  modulated  amplifier  is  modulated 
100  per  cent. 

Tetrodes  and  pentodes  may  be  used  as  modulated  amplifiers,  If  low  distortion  is  re- 
quired, it  is  necessary  to  modulate  the  screen-grid  voltage. 

As  in  driver  modulation,  it  is  usually  not  necessary  to  modulate  the  screen  voltage 
completely.  The  modulating  voltage  may  be  obtained  from  the  plate  circuit  of  the  tube 
through  a  blocking  capacitor  and  dropping  resistor  or,  more  conveniently,  by  a  tertiary 
winding  on  the  modulation  transformer. 

Let  F  =  modulation  factor;  EM  —  maximum  plate  voltage  averaged  over  an  r-f  cycle; 
EC  —  d-c  plate  voltage;  WLP  =  watts  output,  complete  modulation;  WLC  =  watts  out- 
put, normal  carrier;  WM  —  modulator  output  power;  and  Wei  —  modulated  amplifier 
input  power. 

The  following  formulas  show  the  current  voltage  relations  in  a  plate-modulated  amplifier: 

EM  =  Bed  +  F}  (13) 

WLP  -  1.5TFLC  (14) 

,.  (15) 

The  last  equation  assumes  that  the  modulated  amplifier  matches  the  output  impedance 
of  the  modulator  either  directly  or  through  a  transformer,  since,  owing  to  limitation  of 
the  plate  swing  of  the  modulator,  incorrect  matching  may  limit  the  modulation  factor, 
even  though  the  modulator  is  potentially  capable  of  producing  complete  modulation. 

17.  COMPARISON  OF  MODULATION  SYSTEMS 

Any  comparison  of  modulation  systems  develops  into  a  comparison  of  means  of  getting 
a  specified  amount  of  modulated  power  into  a  load,  or  antenna,  under  specified  conditions 
of  operation,  weight,  portability,  fidelity,  etc.  For  light-weight,  portable  transmitters, 
grid  modulation  has  some  advantage,  although  it  is  not  capable  without  feedback  of  high 
fidelity.  For  most  other  applications  plate  modulation  is  used  either  directly,  in  the  last 
r-f  amplifier,  or  in  a  stage  followed  by  one  or  more  r-f  amplifiers.  In  the  last  case,  such 
amplifiers  are  usually  of  the  high-efiiciency  Doherty  type.  For  transmitters  operating  in 
the  standard  broadcast  band  where  a  transmission  is  almost  always  over  a  single  path, 
both  the  "high-level"  system  of  modulation  in  the  last  r-f  amplifier,  and  the  "low-level" 
system  using  Doherty  amplifiers,  are  satisfactory.  However,  where  multipath  transmis- 
sion exists,  and  incidental  phase  modulation  may  cause  objectionable  distortion,  high- 
level  modulation  is  generally  preferred. 

FREQUENCY  MODULATION.  Equation  (4),  in  Article  14  was  shown  to  represent 
the  modulated  r-f  current  for  either  the  frequency-modulated  wave  or  the  phase-modulated 
wave.  It  will  be  noted,  however,  that  the  frequency  swing,  in  frequency  modulation, 


7-76  VACUUM-TUBE  CIRCUIT  ELEMENTS 

>».*.    a*ijM±.m\   £«wk*»s«sjb*'iJ»««r   4m  ff'SVTt*!^    oV ! 


M  given  by: 


»  piw»  abift  IB 
Ktv  m  fi^pwfso*  swing  in  eyelet  per  second. 

I*  »  moduUti&#«igaa2  frequency  in  cycles  per  second. 
Method*  of  placing  fwquency  aad  phase  modulation  axe  discussed  more  fully  in 

8- 


BIBLIOGRAPHY 

CMbv  Morny  G  ,  Carfter  a»4  Sid*  Vraqpmqr  Bslatioaa  with  Mdtitone  Frequency  or  Phase  Modu- 
|«fMHL  JICA  «»^  V«l.  I,  lOi-UW  <*ahr  I««5: 


DETECTORS 

Bj  ycmoa  D.  Laad«ii 

esmootton,  ®r  4«tect^m,  is  the  process  whereby  a  wave  resulting  from  modulation 
w  10  operated  upoa  that  a  wave  is  obtained  having  substantially  the  characteristics  of 


. 

qttftre-v  «aa  10  that  form  01  detection  which  depends  on  the  fact  that  e  of 
the  pow*r  &&TWB  for  &  Bma4iii€$ur  Hrniit  exdted  by  a  modulated  wave  contains  the  dif- 
lcran««  frvqiWMy  between  the  earner  mad  oae  o!  its  sidebands,  which  is  the  original 


Lisear  detect!®®  is  that  form  c£  detection  in  which  the  output  voltage  under  considera- 
IMMI  «  Milwuuilialb*  prop^tiiMial  to  the  EBstant&aeoiis  peak  carrier  voltage  throughout 
tW  wwf  ml  range  @l  the  deteciiiig  de\ioe. 

m  SQUARE-LAW  BETECTIOH 


Amy  d^Tks®  having  a  tKaa-linear  cmrent-roltage  characteristic  will  serve  as  a  detector. 
droet*  feAA  ©tri^ra  or>nlais  a^!  tlt^mkmic  VM^ium  tubes.    (See  Figs.  1T  2,  and 
adwla&ed  voltage  is  of  the  form  given  in  eq,  (5),  p.  7-72,  which,  when 
a^di*>-fre<4uenc>-  terms  o!  the  form 


( 


SQUARE-LAW  DETECTION 


7-77 


i  / 


1.5 


-^.5 


-1.6-ls.2-.8  -.4     0     .4     .8    1^1.6 

Impressed  VtSUarge 

FIG.  1.     Static  Characteristic  of  Iron  Con- 
tact on  Ferro-silicon,  25%  Fe 


1.1 

1.0 

.9 

3-8 
g 
I'7 

1-6 

i-s 

£ 

.3 
.2 

i 

i/ 

\i 

f 

1 

y 

/ 

/ 

/ 

/I 

, 

/ 

n 

^ 

/ 

-10123456 
Bias  Voltage 

FIG.  2.    Static  Characteristic  of  Diode 


rent  In  MTfflampQres 
to  10  G)  &>  •£ 
o  in  o  01  i 

^  I 

1 

I 

1 

j 

1 

/ 

/ 

/ 

0 

EL 

/ 

/ 

/ 

10 

/ 

/ 

/ 

0 

-r^^ 

^ 

—20           -16           -12            -8             -4                 0 
Bias  Voltage 

FIG.  3.    Static  Characteristic  of  Triode 


7-78 


VACUUM-TUBE  CIRCUIT  ELEMENTS 


There  are  two  terms  present  of  each  original  signal  frequency,  resulting  from  the  beat- 
ing of  two  sidebands  against  the  carrier.  From  this  it  is  seen  that  there  will  not  be  detec- 
tion tinlesa  the  carrier  is  preaent,  but  that  the  carrier  and  one  sideband  are  sufficient.  If 
regard  is  paid  to  differences  of  phase,  terras  of  the  same  frequency  may  be  added;  the 
terms  preeent  and  their  phase  angles  are  given  in  Tahle  1. 

Table  JL     Square-law  Detection 

ExprweBkaa  for  ita  AmBo-lreqaency  Current  resulting  from  Square-law  Detection  of  a 
Mid  Bot&  Sid*  Baada.     ^mEar  Terras  Appear  for  each  Pair  of  Audio  Components. 


T«m* 


-f  r" 


Phase  Angle 

Triple-primed  Values  of  Resistance,  etc. 
Apply  to  Original  Modulator  at  Sending  End. 


r  4-  rp 


•  —  tsn~l  - 


•  +  tan- 


•  —  tan'1- 


•  —  fean"1- 


r  -frp 


(2  **»  *  -f  r  ft) 


.  _  taa~^  - 


With  only  <HW  midio  ecmapoaeat  preaent  the  per  cent  modulation  is  2QOEn/Es  =  M. 
t&e  per  cent  se©c^i<i  hmrmoaie  present,  in  terms  of  the  fundamental,  is  Jkf/4.    For 

per  cent  mcwItiifctkMii  this  system  is  therefore  not  satisfactory.  When  more  than 
®m  signal  !rt»qi^ac>'  m  pneront  as  modulation  on  the  carrier  wave,  extraneous  frequencies 
art  produced  ec«rr««^>cm4Mi«  to  UM  sum  and  difference  frequencies  of  all  the  signal  com- 


w|«im«4aw  ^tortw  It  ixwr  Uurgeiy  used  only  for  detection  of  carriers  having  small 
a  and  for  the  reception  of  single  sideband  signals.  It  is  the  onlv 
loam  whkfc  will  wwrk  vith  theee  latter.  In  detecting  single  sideband  and  carrier,  second 
hmm&mm  ol  tfe«  ai^^  coisapo^tiit^  are  not  produced,  but  frequencies  corresponding  to 
tli*  «MHi  aad  dtfftmncwi  feetureea  tte  frecjuencies  of  the  components  of  the  original  signal 
art  prwumt. 

BETECTOE  CIRCUITS.  Any  non-linear  circuit  excited  by  a  voltage 
i  to  that  ii^  «t«tk  ehatraeteristic  over  the  regkm  used  is  closely  a  parabola  is 
m  a  ^wsai^law  <4et««$or.  When  the  vacuum  tube  is  used,  detection  may 


Load 


FKL4. 


r,  Bwtaum  Coup4**i  to  ^HMsccdiog  Ampli&er  Tube 

eiroak®.    Figiire  4  shows  a  circuit  which  is  biased  practi- 
tstfeas  place  owing  to  the  curved  mutual 


LINEAB.  DETECTION 


7-79 


The  voltage  at  any  frequency  k  impressed  on  the  amplifier  tube  is 


ELK 


dep 


(2) 


For  high  detected  voltage  the  external  plate  impedance  at  the  carrier  frequency  (ZB)  should 
be  small,  but  that  at  the  signal  frequencies  (zk)  should  be  large.  The  high  audio  impedance 
is  shunted  by  C,  to  achieve  this  result. 

Figure  5  shows  a  circuit  in  which  detection  occurs  in  the  grid.  The  grid  and  cathode 
elements  act  as  a  diode  detector.  The  a-f  currents,  resulting  from  demodulation,  produce 
an  a-f  voltage  across  the  grid  leak  and  so  between  grid  and  cathode.  This  voltage  controls 
the  electron  stream  so  the  tube  also  acts  as  an  amplifier.  A  high  audio  impedance  and 
low  radio  impedance  are  necessary  so  that  the  grid  leak  is  shunted  by  a  small  condenser. 


Input 


Load 


FIG.  5.    Grid  Current  Detector  (Z>),  Transformer  Coupled  to  Preceding  Tube  (JL) 

This  type  of  detector  broadens  the  tuning  of  the  preceding  tuned  circuit  (owing  to  the  low 
impedance  of  the  grid  when  positive),  and  overloads  badly  on  strong  signals,  owing  to  the 
r-f  and  d-c  components  of  the  plate  current.  It  is  very  sensitive,  however. 


19.  LIKEAR  DETECTION 

When  very  large  signals  are  applied  across  non-linear  circuits  the  power  series  con- 
vergence based  on  the  static  characteristics  is  so  slow  that  the  method  is  not  useful.  The 
equation  for  the  modulated  wave  can  optionally  be  written  in  the  form  (see  eq.  6,  p.  7-72) : 


-  2  2     ^  cos  (uKt  +  0*)     cos  (wrf  -f-  6,) 


(3) 


which  represents  a  voltage,  of  fixed  frequency,  whose  amplitude  is  slowly  varying  about  a 
mean  value  Es.  The  sum  (S)  within  the  brackets  will  represent  a  voltage  of  the  same 
wave  shape  as  the  original  modulating  voltage,  provided  the  modulation  and  transmission 
have  been  distortionless.  The  instantaneous  peak  voltage  of  the  carrier  current  is 


-22]T 

KES 


+ 


(4) 


Rectification  diagrams  of  the  detecting  device  are  obtained  by  plotting  the  total  direct 
current  in  the  detector  circuit  against  various  steady  values  of  rms  (or  peak)  carrier  voltage. 
(See  Fig.  6.)  Then  when  Em  is  varied  (in  accordance  with  the  signal  voltage)  this  current 
varies;  this  detected  current  is  equivalent  to  a  steady  direct  current,  wbose  value  is  de- 
termined by  the  value  of  direct  current  which  flows  when  Em  =  Ett  on  which  is  superposed 
a  varying  current  which  will  follow  the  variations  in  JEm  (and  so  in  the  signal  voltage)  more 
or  less  closely,  depending  on  the  linearity  of  the  rectification  diagram.  Since  the  periodic- 
ity (&>«)  of  the  carrier  voltage  is  not  of  importance  except  in  fixing  the  circuit  impedances, 
the  rectification  diagrams  can  be  obtained  by  using  any  convenient  frequency,  such  as 
60  cycles,  rather  than  an  actual  carrier  frequency,  provided  the  circuit  impedances  are 
made  equal  in  the  two  cases. 


YACUUH-TUBE  CIRCUIT  ELEMENTS 


It  will  b©  noticed  from  tit*  rectification  diagrams  shown  in  Figs.  7-12  that  the  incre- 
ment in  direct  current  due  to  a  varying  grid  voltage  varies  approximately  as  the  square  of 
the  maximum  value  of  this  voltage  when  Em  is  small  (as  required  by  the  power  series 

expansion),  but  that,  as  Em  increases,  the 
d-c  response  varies  linearly  with  Em. 
When  it  does  so  the  distortion  introduced 
in  detection  will  be  small. 

As  an  illustration,  consider  the  plate- 
current  detector  shown  in  Fig.  4.  Currents 
of  radio  frequency  are  by-passed  by  C,; 
hence  the  only  current  which  flows  through 
r  when  Em  is  steady  is  a  direct  current. 
If  the  value  of  this  current  were  varied  in 
accordance  with  the  signal  voltage,  the 
varying  voltage  impressed  on  the  grid  of 
the  succeeding  amplifier  tube  would  be 
simply  the  signal  voltage.  If  the  variation 
of  direct  current  against  Em  were  given  by 
the  curve  of  Fig.  6  and  the  average  value 
of  Em  were  equal  to  OA,  then  the  varia- 
tktn  0f  rumemt  would  b©  exactly  proportional  to  the  signal  voltage,  even  for  100  per  cent 
mGdulotum,  provided  Em  <  OB.  This  njeans  tiiat  there  would  be  no  distortion  introduced 
by  the  defector. 

Although  ao  drrk©  avmHa&kf  at  present  has  exactly  the  characteristic  of  Fig.  6,  certain 
eiyetale  appnminiAfe©  it,  aud  the  vacuum  tube  can  be  made  to  approach  it  by  a  proper 


<u 

I 

C 
Fw 

x 

/ 

Et 

-*.< 

«»«.., 

XI 

r- 

X 

/f 

>                   A                    B 
,  C     Ideal  Defcwtor  Ch»r»rleii8tk  (Bectifica- 

A  4fio4» 


*ctifi«*tian  Da«f»m  for  E^xfe  Detector 

.    f%nr«  7  sfeows  a  set  of  curves  f  or  a  diode  f  torn 
<ij»&mm  fer  aaay  load  nssfeiaaee  can  be  ofc^ained;  the  diagram  for 
m  rnmem  m  F%.  8,  wfekii  «o«iM  also  have  been  obtained  expeiimentallv 
t  h*  aeftrebt  apt^^di  to  liaaar  &s*e«$i©B  of  anj-  kaown  detector. 


LINEAR  DETECTION 


7-81 


When  the  characteristic  is  not  a  straight  line,  a  Taylor's  series  expansion  can  be  used 
in  terms  of  epa  and  the  change  (em)  hi  Em: 


dip 


dip 


(5) 


and  em,  tp  is  the  plate-current  ordinate  of 


80 


50 


s 

40 


230 


20 


10 


where  ipc  is  the  current  due  to  the  variations  ej 
the  rectification  diagram,  and  epa  is  the 
plate  potential  due  to  ipa  and  em.  The 
reciprocal  of  dip/dep  is  the  analog  of  the 
plate  resistance  used  in  Section  5,  and 
has  been  named  the  detection  plate  resist- 
ance. The  derivative  dip/dem  is  the  slope 
of  the  rectification  curve — it  is  analogous 
to  the  mutual  conductance  of  a  triode  and 
is  called,  the  detection  mutual  conductance 
or  the  transrectification  factor. 

All  the  calculations  given  for  class  A 
amplifiers  now  may  be  utilized  in  the 
detector  circuit,  including  calculations  of 
distortion,  detection  efficiency,  etc. 

LINEAR  DETECTORS.  As  stated 
above,  the  diode  gives  the  nearest  ap- 
proach to  linear  detection  known  today. 
It  has  the  objection  of  putting  a  fairly 
small  resistance  in  parallel  with  a  tuned 
circuit  and  so  considerably  broadening 
the  tuning.  When  a  diode  is  used  the 
applied  r-f  voltage  should  be  made  as  high 
as  the  tube  can  stand  and  a  load  resistance 
selected  to  give  minimum  distortion. 

The  triode  is  not  as  distortionless  as 
the  diode  but  sometimes  fits  in  better 
with  the  circuit  requirements.  Rectifica- 
tion diagrams  of  a  triode  alone  and  with 
load  are  given  in  Figs.  9  and  10.  In  gen- 
eral, the  plate  voltage  should  be  as  high  as  possible,  limited  only  by  the  supply  available 
or  the  limits  of  the  tube  itself  (flashover  or  heating). 

When  the  plate  voltage  has  been  determuied,  rectification  diagrams  of  the  tube  alone  at 
various  grid  biases  are  obtained.  If  it  is  desirable,  as  is  usual,  to  use  the  tube  with  carrier 
voltages  which  are  100  per  cent  modulated,  the  plate  current  should  not  reduce  to  zero  for 
any  finite  value  of  Em,  or  part  of  the  modulation  will  produce  no  effect.  In  other  words, 
the  tube  should  not  be  biased  below  cutoff.  The  best  results  will  be  obtained  if  the  tube 

is  biased  at,  or  slightly  above,  cutoff  (on 
the  static  characteristic);  this  condition 
will  be  indicated  on  the  rectification  dia- 
gram by  a  small  plate  current  for  Em  =  0. 
For  the  tube  of  Fig.  9  the  grid  voltage  is 
thus  determined  as  either  —1.5  or  —3.0 
volts. 

To  determine  the  proper  load  resist- 
ance, rectification  diagrams  of  the  tube 
with  different  loads  (see  Fig.  10)  are  ob- 
tained. If  Em  is  varied  over  a  straight 
portion  of  one  of  these  curves,  distortion- 
free  detection  will  result;  if  the  variation 
is  over  a  curved  portion  the  distortion  may 


0  5  10  15  20  25  30 

RMS   Signal  Input  Voltage 

FIG.  8.    Load    Rectification    Diagram    for    Diode 
Using  500,000-ohm  Load 


6       8      10     12 
Eja.  (S-M-S.)  in  Volts 


FIG.  9.    Rectification     Diagrams  of  Triode  (Alone). 
All  data  were  taken  with  6Q-cycle  impressed  voltage. 


be  calculated  as  for  an  amplifier.  (See 
p.  7-07.)  For  the  tube  used  here  a  load 
impedance  of  30,000  ohms  would  cause 
the  tube  to  introduce  a  very  low  percentage  of  distortion  for  a  carrier  having  an  average 
rms  value  of  6.25  volts,  and  100  per  cent  modulated.  The  difference  between  curves  C 
and  E  illustrates  the  gain  in  efficiency,  but  increased  distortion,  obtained  by  by-passing 
the  load  resistance. 

When  the  triode  is  used  as  a  detector  with  large  signal  voltages  of  high  percentage  of 
modulation,  operation  with  niinimum  distortion  is  always  accompanied  by  a  positive  grid 


7-S2 


VACi'UM-TUBB  CIRCUIT  ELEMENTS 


fwiniE  aad  *o  a  flow  o(  grid  current.    This  causes  a  low  input  impedance  and  so  decreases 
the  **<ertivi?y  of  the  tuned  circuit  just  preceding  the  detector. 

Th*  w  of  *  tetrode  <**e  Fig.  ID  makes  it  possible  to  obtain  rectification  diagrams 
wiirh  r*a*h  ^turation  for  negative  v&Jue®  of  control  grid  voltage,  so  giving  straighter 
operating  run**  without  lowering  the  input  impedance  of  the  tube.  The  same  method  of 
foikwed,  the  only  additiaaal  factor  being  the  proper  voltage  to  use  for  the 
_ _ ^  screen  grid.  The  method  employed  in  se- 

-jUa 


.gfof ... 


lecting  the  screen-grid  voltage  depends  to  a 
large  extent  upon  the  type  of  plate  load. 
In  general,  a  higher  screen-grid  voltage  gives 
a  better  rectification  diagram  (straighter 
and  steeper).  However,  when  there  is  a 
large  d-c  resistance  in  the  plate  circuit,  the 
plate  voltage  falls  considerably  for  large 
signal  voltages,  so  that,  if  the  screen-grid 
voltage  is  comparable  with  the  plate  volt- 
age, electrons  will  land  on  the  screen  grid 
rather  than  on  the  plate. 

When  reactance  coupling  is  used  the 
screen-grid  voltage  should  be  the  maximum 
consistent  with  the  plate  voltage.  For  this 
type  oJ  coupling  the  d-c  resistance  of  the 
detector  plate  load  is  small  enough  so  that 
large  carrier  voltages  will  not  cause  sufficient 
drop  for  tht  plate  voltage  to  beeome  comparable  to  the  screen-grid  voltage.  Therefore, 
there  w  ao  Beeeerifcy  ©£  keepsf  Uw  acne^n  voltage  low  to  preve&t  saturation  (on  the 
diagram),  &nd  il&e  gala  in  mutual  conductance  with  high  screen  voltage  raises 
,uvh>  md  maximum  cmtpm  as  well  as  decreases  the  curvature  of  the  rectification 

is  iaa©d  the-  selection  of  the  optimum  screen-grid  voltage  is 
>  probbm  la  to  select  a  value  of  screen-grid  voltage  that  will 
output  at  the  normal  values  of  input  and  yet  not  cause  over- 
of  input  voltage.    Two  rectification  diagrams  for  different 
a  taigti  resistance  bad  are  shown  in  Fig.  12.    The  curve  for 

£ 


lilt 


WVra  rewtiaiew 


£wn»iimfcl® 

for  the  larger 


«  7S  Tc4t«  u 

tub©  ovwrfewli  for 


-  —    12.     Load   HjsctlficatioD   Diagrams    of   a 

Tetrode,    Slewing   t^e   Elect  of  Screen   Grid 

Voltage  oe  Satomtioft.    (Resistance  load.) 

«  40  volts  for  inputs  up  to  a  point  A,  but  the 
•*«  -4.  It  is  thtis  necessary  to  know  the  maximum 
os  the  detector,  and  the  screen-grid  voltage  which 


BIBLIOGRAPHY 


)  Voha^a-^PU^  R«ctifimtioa  ni^  the  ^^-vscniim  TAxte. 


VACUUM-TUBE  OSCILLATORS  7-83 

OSCILLATORS 

By  Carl  C.  Chambers 

An  oscillator  is  a  non-rotating  device  for  producing  alternating  current,  the  output 
frequency  of  which  is  determined  by  the  characteristics  of  the  device.  Oscillators  can 
be  conveniently  divided  into  three  classifications:  (1)  vacuum-tube  oscillators,  (2)  elec- 
tromechanically  controlled  oscillators,  and  (3)  spark-gap  oscillators. 

20.  VACUUM-TUBE  OSCILLATORS 

The  most  generally  used  oscillator  in  the  field  of  communication  engineering  is  the 
vacuum-tube  oscillator.  Since  a  vacuum  tube  can  act  as  an  amplifier  of  power,  it  can 
act  as  an  oscillator.  The  power  needed  in  the  grid  circuit  can  be  supplied  by  the  plate 
circuit.  A  vacuum-tube  oscillator  circuit  acts  as  a  power  converter,  changing  d-c  power 
into  a-c  power,  having  a  frequency  determined  by  the  parameters  of  the  circuit.  The 
efficiency  of  vacuum-tube  oscillators  can  be  above  90  per  cent,  although  for  many 
purposes  they  operate  at  low  efficiency.  Vacuum  tubes  are  constructed  to  dissipate  as 
high  as  125,000  watts,  the  power  output  of  oscillators  using  such  tubes  being  much 
greater. 

SIMPLE  OSCILLATOR  CIRCUITS.  In  ordinary  vacuum-tube  oscillators,  power  is 
fed  into  the  grid  circuit  from  the  plate  circuit  by  means  of  either  electrostatic  or  electro- 
magnetic coupling  between  these  circuits.  When  sufficient  voltage  of  proper  phase  is 
introduced  into  the  grid  circuit,  the  a-c  component  of  the  plate  current  will  persist,  or,  in 
other  words,  the  circuit  wiU  oscillate. 

Many  oscillator  circuits  have  been  devised,  several  of  which  are  shown  in  Fig.  1.  In 
each  of  these  circuits,  the  oscillatory  circuit  is  the  mesh  containing  L  and  C,  In  the 
Hartley  circuit,  the  inductance  L  is  tapped  (not  necessarily  at  the  center) ;  in  the  Colpitts 
circuit,  the  capacitance  C  is  the  series  capacitance  of  Ci  and  C%.  The  tuned-plate  tuned- 
grid  oscillator  has  two  separate  oscillatory  circuits,  one  in  the  plate  circuit  and  one  in  the 
grid  circuit,  both  tuned  to  approximately  the  same  frequency.  In  each  of  these  oscillators 
the  load  is  coupled  to  the  oscillatory  circuit.  This  coupling  is  usually  but  not  always  in- 
ductive coupling. 

The  voltage  introduced  into  the  grid  circuit  due  to  the  plate  current,  in  (a)  and  (6),  is  the 
drop  across  part  of  the  oscillatory  circuit.  In  (c),  (e),  and  (/),  it  is  the  voltage  across  one 
winding  of  the  transformer,  one  winding  of  which  is  the  inductance,  I/,  of  the  oscillatory 
circuit.  And  in  (d)  it  is  the  voltage  drop  across  the  grid  oscillatory  circuit  due  to  the 
current  through  it  and  the  grid  to  plate  capacitance.  The  peak  value  of  this  feedback 
voltage  in  each  case  must  be  great  enough  to  cause  oscillation  to  persist.  For  stable 
operation,  it  should  be  roughly  two  and  one-half  to  three  and  one-half  times  the  d-c  grid 
voltage  necessary  to  bias  the  tube  to  plate  current  cutoff  at  the  operating  d-c  plate  poten- 
tial. 

The  capacitances,  C.B,  are  by-pass  condensers,  and  the  inductances,  LB-,  are  choke 
coils  arranged  to  prevent  the  alternating  current  from  passing  through  the  d-c  supply 
source,  SB,  or  through  the  biasing  resistor  or  grid  leak,  TB-  The  grid  voltage  in  most 
oscillators  goes  positive  for  part  of  the  cycle.  Thus  the  grid  bias  voltage  is  conveniently 
obtained  by  means  of  a  grid  leak  and  condenser.  This  method  of  biasing  the  oscillator 
tube  tends  to  self-adjust  the  grid  bias  to  permit  operation  over  a  wide  range  of  values  of 
the  feedback  voltage,  since  with  greater  excitation  the  grid  tends  to  go  more  positive  and 
consequently  the  grid  bias  becomes  greater,  resulting  in  essentially  the  same  plate-current 
flow.  Some  oscillators  have  a  comparatively  small  external  grid  bias  (or  self-bias  by 
means  of  a  cathode  resistor)  in  addition  to  the  grid  leak.  This  externally  supplied  bias 
is  a  safety  device  to  prevent  too  much  plate  current  from  flowing  if  the  oscillations  should 
cease  for  any  reason, 

WOK-LINEAR  THEORY  OF  OSCILLATIONS.  "'The  complete  solution  of  even  the 
simplest  oscillator  circuits  is  extremely  complex.  However,  van  der  Pol,  by  means  of  very 
radical  assumptions,  has  obtained  the  solution  of  several  oscillator  circuits.  One  of  these 
solutions  is  for  the  circuit  shown  in  Fig.  I/,  The  following  assumptions  are  made  con- 
cerning the  circuit:  (1)  that  the  grid  bias  voltage  is  obtained  by  means  of  an  external 
electromotive  force;  (2)  that  no  current  flows  in  the  grid  circuit;  (3)  that  the  capacitances 
CB  are  so  large  that  their  a-c  impedance  is  zero;  and  (4)  that  p,  the  amplification  factor, 
is  constant.  These  approximations  make  the  solution  useful  only  from  a  qualitative  stand- 
point. 


7-84 


VACUUM-TUBE  CIECUIT  ELEMENTS 


Thf  OMMMiy  condition  for  omllation  to  persist  is  found  to  be  that  a,  the  damping 
eooauat  of  the  rirruit  taken  as  a  whole,  is  negative.  To  a  rough  approximation  the  ampli- 
ttxfo  01  tin*  oftcillation  sa  proportional  to  the  square  root  of  the  ratio  of  a  to  the  coefficient 
of  th*  rob*  t*rm  in  the  power  series  expansion  of  the  characteristic  of  the  vacuum  tube. 

Tb»  WTO  form  of  th®  o*ciU»tion  is  dependent  upon  €,  the  product  of  a  and  LC.  The 
f  UMI  circuit  to  semral  values  oi  «  as  calculated  on  the  differential  analyzer  at 


ijtj 


/-i 


n 


^:C 


•t 


B 

'h 


1.     C<mvim®@&»l  O^fllatw  Circuits 


Moort  Sebool  c^  Electric*!  En^iuwring  are  stown  in  Fig.  2.    For  smaH  values  of  e, 


tht 


ti*l!y  mfii«sidaat  the  frequency  being  given  by 


For 


f  t,  the  o»cifa^s9ti«  d@p^i  ra<H«aUy  from  sme  waves.  This  form  of  oscillation 
«*IM  rdi!t*®feia  ««aU«ltMi.  It  oerurs  ia  the  multivibrator,  the  human  heart,  the 
ie  ram,  etc-  Van  der  Pol  pws  the  period  f  or  «  i?>  1  as  roughly  equal  to  2e. 
GOWDITIOirS  FOR  S!I#~0SCttLATIOlf.  For  smaM  amplitudes  of  oscillation,  the 
B$»4iia«fcr  rm®i&$a&&m  ©f  UMI  |slml«f  SMK!  grid  circuits  can  be  approximated  by  linear  resist- 
aaa^e^  Tfe®  ©e»ditkm  for  ilsw©  §maH  o*ciU»tktti8  t©  build  up  is  that  the  damping  constant 
&  Fw$*tive.  Tlas  u  th«  fsmwlitw^  for  unatabb  systems  in  traa^ent  circmt  theory.  How- 
««».  as  th*  ©neiaatti^M  meimw  in  amplitude,  the  linear  circuit  theory  fails. 

U»*Ptip»  h*^  ^@wa  that.  um«irr  ^aady-«tai«  operatioii  for  any  freqijency  component 
«f  A®  funw&l*  &  m^*toe«r  r®^^a»cft  ew  be  replaced  by  an  e*j*iivaieBt  impedance  which 
I^IF  a^wosimattd  by  a  pure  nssistanee.    When  the  circuit  oscillates 
.  ti»  HMMAMOMHUI  ir««jntacy  ot  oi^iat^i  has  a  coolant  amplitude  so  that  the 


VACUUM-TUBE  OSCILLATORS 


7-85 


plate  to  filament  circuit  in  the  tube  can  be  replaced  by  such  an  effective  resistance.  In 
order  for  steady-state  currents  of  a  given  frequency  to  flow  in  a  circuit  containing  no  emf 
of  that  frequency,  the  determinant  of  the  coefficients  of  the  currents  in  the  mesh  equations 
must  be  zero.  An  oscillator  must  satisfy  such  a  condition.  The  approximate  resulting 
conditions  are  noted  for  most  of  the  circuits  in  Fig.  1.  The  value  of  the  effective  plate 
resistance,  rp  ,for  any  amplitude  is  given  approximately  by  the  inverse  slope  of  the  secanc 
joining  the  limiting  points  of  operation  on  the  plate  characteristic  of  the  tube.  By  the  ' 
reverse  calculation  the  magnitude  of  oscillation  for  any  given  circuit  can  be  predicted. 


6=0.1 

r\ 

^-v^ 

/~\ 

/ 

\      / 

\ 

X 

_/ 

\y 

\  / 

\     , 

^ 

M 

10 


4- 
0 

I 

_/~\ 

/  \ 

I 

\  /! 

V 

j 

\  I 

\  / 

i 

^ 

v 

20 


25 


10 


15 


20 


/    \ 

e=i.o/ 

\ 

/  \ 

y\ 

/         ^ 

,    / 

\ 

/   \ 

\ 

/ 

A  / 

\  y 

r        v 

\ 

y 

\/ 

/ 

3               5               10  ~         15         w   20             2 
-ft 

/N 

/  ^e=2.o 

!  X 

• 

/      \ 

/ 

\ 

1      \ 

\ 

/ 

I  / 

\ 

/ 

\  i/ 

\/ 

\ 
1 

/ 

0                5               10              15              20             2! 
-H 

PIG.  2.     Solutions  of  van  der  Pol's  Equation  for  the  Non-linear  Theory  of  Oscillations: 
(fl  ~  «<1  ~  *2>  |  +  u  -  °) 

CURRENT  AND  VOLTAGE  RELATIONS  IN  SIMPLE  OSCILLATOR  CIRCUITS.  , 

As  stated  at  the  beginning  of  this  chapter,  the  vacuum-tube  oscillator  is  simply  an  ampli- 
fier arranged  so  that  the  power  needed  in  the  grid  circuit  is  supplied  by  the  plate  circuit. 
Ordinarily,  the  amplifier  so  arranged  belongs  to  that  group  known  as  class  C  amplifiers. 

The  current  and  voltage  relations  of  a  typical  oscillator  are  shown  in  Fig.  3  for  one 
complete  cycle.  The  plate  current  flows  during  the  portion  of  the  cycle  when  the  instan- 
taneous plate  voltage  is  least  so  that,  since  the  energy-  dissipated  at  the  plate  of  the  vac- 
uum tube  is  the  integral  of  the  instantaneous  plate  voltage  times  the  instantaneous  plate 
current,  the  tube  plate  loss  is  least  when  the  current  flows  for  as  small  a  portion  of  the  cycle 
as  possible.  When  plate  current  flows,  the  grid  voltage  is  positive  with  respect  to  the  grid 
bias  necessary  for  plate-current  cutoff.  Thus  the  duration  of  plate  current  decreases  and 
consequently  the  plate  efficiency  increases  with  simultaneous  increases  in  the  grid  bias 
and  the  a-c  grid  voltage.  However,  increasing  the  grid  voltages  in  this  way  increases  the 
losses  in  the  grid  circuit.  The  a-c  grid  voltage  usually  has  a  peak  value  of  two  and  one- 
half  to  three  and  one-half  times  the  grid  bias  necessary  for  plate  current  cutoff  at  the  d-c 
plate  voltage. 

Since  plate  current  flows  only  during  minimum  instantaneous  plate  voltage,  the  mini- 
mum plate  voltage  is  of  major  importance.  The  impedance  of  the  oscillatory  circuit  at 
resonance  is  adjusted  so  that,  for  the  normal  power  output,  the  peak  a-c  voltage  across  it 
is  slightly  smaller  than  the  applied  d-c  plate  voltage.  Therefore,  the  plate  voltage  causing 
current  to  flow  is  small  compared  with  the  d-c  plate  voltage,  since  it  is  essentially  the  d-c 
plate  voltage  minus  the  peak  a-c  voltage  across  the  oscillatory  circuit.  Thus  the  a-c  and 
d-c  voltages  remain  practically  equal  as  the  d-c  voltage  is  varied.  It  is  this  equality 
between  the  d-c  and  a-c  voltages  that  makes  the  plate  modulation  of  oscillators  and  class 
C  amplifiers  so  nearly  linear. 

In  ordinary  triodes,  the  maximum  plate  current  flows  for  a  positive  value  of  the  grid 
voltage  somewhat  less  than  the  rrdninmm  instantaneous  plate  voltage.  In  most  oscillators 
at  full  load,  the  relation  between  the  plate  voltage  and  the  grid  voltage  is  such  that  the 


7-S6 


VACUUM-TUBE  CIECtJIT  ELEMENTS 


YalW  Ol    in«  IP***  i«aut  MS  ueoi.  M^W*****"^^   «j    — _  -i* 

Tolt*«e  is  about  0.8  of  the  minimum  instantaneous  plate  voltage,     in  small 

Zero  Time 


CWTWB*  Km!  Vota*  EelstaQiaa  of  a  Representative  OseSJator 


>  dficieacy  is  so*  Tital  the  value  ©I  tl*e  grid  leak  can  vary  over  wide  limits;  a  value 
taw  from  &  few  thousand  ©hms  to  a  megohm  may  be  determined  by  trial  and  error 
until  the  oB«aEator  operate®  satisfactorily. 

Sou»tii3i©8  wl»n  the  time  e<»^«at  ®f  the  grid  leak  sod  condenser  is  high  and  when 
Oit  rati©  of  the  »HD  pU^a  voatape  to  th«  a-c  g?id  voltage  is  low,  "blocking"  will  take  place. 
""  "  !  *&  a  r»Iax»tkBa  oscallatioa  and  is  due  to  tfee  grid  being  at  a  higher  potential  than 
»  a©  that  tbe  seoo^ary  ^a»k»i  I ttsm  Hi©  grid  causes  the  grid  leak  and  condenser 
p  a  iugh  b$&i  wfekli  stop*  tii«  o®cilliiti<»  imOi  the  charge  leaks  off  the  condenser, 
at  whieh  «inw  «^il»ti«^i  y^m  atari  «ad  the  cy^»  it^>eats  itself .  It  can  be  corrected  by 
i&mttiixte  th*  f»lk>  ol  ta*  *-e  plate  voita^  to  tfee  M  grid  voltage. 

0BCP<lnJt.TOaTY  CIEC0IT  DESIGN.    The  heart  <^  any  sinusoidal  oscillator  is  the 
„  ol  aa  inductance  and  a  capacitance  oonnected  in  series. 
h  ^ai  mspplaed  to  this  circuit  from  &  d-c  source  by  a^ans  of  a  vaeuum  tube,  and 
11  m  i*ken  «nw^r  Inoaa  it  by  co^plkig  tlae  load  to  it  in  otae  of  varioiis  ways.     The 
swaH**-  to  A»t  ol  the  Sywfeeel  im  a  nsdproci^iiig  steam 


VACUUM-TUBE  OSCILLATORS 


7-87 


The  oscillatory  circuit  must  store  the  energy  used  by  the  load  long  enough  so  that  it  can 
supply  the  load  continuously  even  though  it  receives  energy  only  during  a  small  portion 
of  the  cycle.  It  has  been  found  empirically  that  in  order  to  do  this  job  effectively,  i.e., 
to  give  a  good  wave  form,  the  ratio  of  the  peak  energy  stored  per  cycle  must  be  at  least 
twice  the  energy  fed  into  the  load  per  cycle.  From  ordinary  circuit  theory  it  is  known 
that  a  load  resistance  paralleling  a  tuned  circuit,  or  coupled  to  a  tuned  circuit,  can  be 
replaced  approximately  by  a  resistance  in  series  with  the  tuned  mesh.  When  this  is  done 
the  ratio  of  the  peak  stored  energy  to  the  energy  dissipated  per  cycle  is 


VI 


r-r 


(1) 


where  V  is  the  rms  voltage  across  the  capacitance,  C,  and  across  the  inductance,  L;  I  is 
the  rms  current  through  the  inductance  and  capacitance;  r(=  L/rzC,  where  TL  is  the 
shunting  resistance  load)  is  the  effective  series  resistance  of  the  circuit,  assumed  to  be 
small  compared  with  2x  times  the  frequency,  /,  times  L.  Thus  if  the  ratio  is  to  be  greater 
than  2,  27r/L/r(=:  Q)  must  be  greater  than  4x.  This  Q  refers  to  the  inductor  together 
with  the  equivalent  resistance  due  to  the  load. 

On  the  other  hand,  if  the  ratio  L/r(—  TL€  =  TL/^fiL)  is  made  too  large  the  resonant 
resistance  of  the  circuit  will  be  too  high  to  obtain  a  satisfactory  power  output.  It  is 
therefore  necessary  to  compromise  between  power  output  and  wave  form.  It  is  in  gen- 
eral good  practice  to  make  the  ratio  of  the  stored  power  to  the  power  output  about  2, 
i.e.,  to  make  Q  =  4x.  L  and  C  are  then  given  in  terms  of  the  power,  the  frequency,  and 
the  voltage  (essentially  the  d-c  plate  voltage)  as 


L 
L 


(2) 


c=w  (3) 

These  formulas  are  not  meant  to  be  critical.  If  good  wave  form  is  more  important,  use 
a  value  of  P  in  these  formulas  greater  than  the  actual  power;  if  power  output  is  more 
important  use  a  value  of  P  less  than  the  actual  power.  This  principle  is  frequently  stated 
by  saying  that  the  wave  form  is  improved  by  decreasing  the  L  to  C  ratio  of  a  tuned  circuit. 
These  circuit  considerations  apply  equally  well  to  the  tuned  circuit  of  a  class  C  amplifier. 

CONSTANT-FREQUENCY  OSCILLATORS.  Using  the  equivalent  impedance  dis- 
cussed under  Conditions  for  Self-oscillation,  above,  (a)  and  (6)  of  Fig.  1  are  special  cases 
of  the  equivalent  circuit  shown  in  Fig.  4.  The  ordinary  mesh  equations  for  this  circuit  are 

-  Ipte  4-  2m)   +  lite  4-  *2  4-  Z*  +  2*m)    -  /ife  +  *»)   =   0  (46) 

•  2«)  4-  Ig(zg  4-  22  4  z*)  =0  (4c) 


Most  of  these  impedances  are  functions  of  the  frequency.    And,  since  the  condition  that 
any  of  these  currents  can  be  other  than  zero  is  that  the  determinant  of  the  coefficients  of 
the  Ps  is  zero,  the  frequency  of  oscillation  is  the 
frequency  which  will  make  that  determinant  zero. 
Thus  the  frequency  depends  not  only  upon  the 
parameters  of  the  oscillatory  circuit  but  also  on  the 
other  parameters  of  the  circuit.    In  order  that  the 
frequency  shall  remain,  constant,  all  these  param- 
eters must  in  general  remain  constant. 

To  keep  the  inductances,  capacitances,  and  re- 
sistances external  to  the  tube  constant  is  a  prob- 
lem in  the  design  and  temperature  control  of  those 
parts.  On  the  other  hand,  regardless  of  the  de- 
sign, the  effective  impedance  of  the  tube  itself 
changes  with  use  and  supply  voltages.  Several 
methods  have  been  devised  to  maintain  essen- 
tially constant  supply  voltages,  and  seasoned  tubes 
tend  to  reduce  the  changes  in  the  tubes  themselves. 
However,  by  adjusting  the  other  circuit  parameters  the  dependence  of  the  frequency  on 
the  tube  impedances  can  be  minimized. 

The  most  common  method  is  simply  to  increase  the  sharpness  of  the  oscillatory  circuit, 
that  is,  decrease  the  decrement  of  the  circuit.  This  is  carried  to  the  extreme  by  the  use 
of  mechanical  resonators  such  as  the  quartz  crystal  and  the  magnetostriction  rod. 


FIG.  4.    The  Equivalent  Circuit  of  Most 
of  the  Circuits  of  Fig.  1 


7-«8 


VACUUM-TUBE  CIRCTJIT  ELEMENTS 


Uewrfyn  baa  d*m>ver*d  that,  fey  proper  adjustment  of  the  impedances  Z<  and  Z&  it 
hb  tn  the  limiting  caae  of  no  harmonic  currents  in  the  tube  to  ehmmate  the  depend- 
of  the  frequency  of  oscillation  on  the  tube  impedances.    Although  this  limiting  case 
new  re&cimi  the  w  of  tlie«  impedance®  tends  to  stabilize  the  frequency.    In  addi- 
tion it  is  sometimes  possible  to  change  the  frequency  by  a, 
simple  change  in  the  parameters  of  the  oscillatory  circuit 
witbowt  disrupting  this  independence  of  the  frequency  on  the 
tube  impedances.    A  circuit  derived  from  the  Colpitts  cir- 
cuit for  this  purpose  is  shown  in  Fig.  5,  where  1/4  —  LzCi/Cz. 
Ttiu«  the  frequency  can  be  adjusted  by  changing  Ci  and  Cg 
proportionally  without  changing  the  calculated  value  of  L±. 
THE    HARMONIC    CONTENT    OF     OSCILLATORS. 
From  the  non-linear  theory  of  oscillators  (see  article  20)  it 
is  known  that  an  oscillator  is  stable  only  if  it  is  non-linear. 
Thus  an  oscillator  always  produces  harmonics  in  its  output. 
Tbe  percentage  of  the  harmonics  becomes  small  when  €  be- 
comes small,  that  is,  when  the  feedback  from  the  plate 

to  the  grid  dretilt  is  small.  A  circuit  applicable  at  low  frequencies  in  which  the 
f«dhaek  can  be  controlled  is  shown  in  Fig.  6.  The  variable  resistance  r  controls  the 
amount  of  ftfxilmek.  Tfam  circuit  not  only  has  a  controlled  harmonic  content,  but  it  also 
is  wry  *tmU«  with  frequency.  It  is  exeeUeat  for  a  laboratory  oscillator  where  r  can  always 
fee  mdiust^d  to  emnpenMte  for  changes  in  tbe  tube  characteristic. 

In  large  oscillators  aad  class  C  arapMSfcrs  a  aeries  tuned  circuit  is  sometimes  connected 
m  aeriee  with  tfa«  plate  eimiit  to  offer  a  high  impedance  to  the  harmonics.  Such  a  tuned 
euwutt,  m  additwa  to  decr^ast&f  the  Ifcrnrsaoaie  content  to  a  marked  extent,  increases  the 


Fi*      5,     A     r<*w«iiit 

qwwry  C»mnt    Pwjwdi   from 
a  C«jfft« 


SEPARATELY  EXCITED  OSCILLATORS,  Separately  excited  oscillators  are  a  special 
C  mmpJi^ers,  The  volt&gp©  an&d  current  relations  of  the  true  oscillator  are 
of  a  properly  operated  daas  C  amplifier.  If,  however,  the  frequency  of 
vt»K«0»  differs  from  timt  at  whicfe  tbe  tube  would  oscillate,  the  plate-current 
wmv®  form  n  M34  aymtiMtnail  about  the  icro  ordinate  shown  in  Fig.  3.  Increasing  the 
diMormaMttry  <rf  tliia  wmv©  by  vmryiag  the  capacitanae  of  the  oscillatory  circuit  increases 
ttet  awrmfe  or  d-r  euirrt^t.  Thus  tlie  freqtiency  of  the  c^cillatory  circuit  is  adjusted  in 
ihitt  t>"pe  of  da^  C  ampliiers  until  the  d-c  ctirrent  is  a  minimum. 

Most  of  the  theory  wad  empirical  relatkus  given  for  oscillators  can  be  applied  to  class  C 


I — <wWwv — j 


OF  OSCILLATORS.  When  two  oscillators  oscillating  at 
freqwiaesfcs  are  loosely  coupled  together,  they  mutually  distort  the  voltage 
acTcw  their  oscillatory  circuits.  Tfeis  distortion  in  turn  distorts  the  current  and 
discussed  above.  This  distortion  causes  a  magnified  distortion  of  the 
plat*  etsrrtat  sine*  rum  a  small  per  cent  change  in  the  oscillatory  circuit  voltage  has  a 
great  effect  o®  th*  minimum  plate  voltage.  This  change 
i  an  addition*]  change  in  the  voltage 
;  the  cwetll&tecy  cirt-uit  drop  directly  ami  indi- 
?  throiigh  th®  «$iaii||e  m  Thus  the  volt- 

aic due  to  lib©  one  oscillator  causes  &  magnifiecl  change  in 
the  other  oecaUator.  Thin  distortion  tends  to  shifi  the 
phaa«»  ©f  the  pl»fce  wit*«©  with  iwpeet  to  the  grid  voltage, 
When  th»  ^iase  ehtft  exc^wls  tfee  limit  alkmmHe  for 
•  omUatiofj,  the  o»«allator  jump*  into  oscillation  at 

h  of  the  other  oa^cilliitcM1. 
Hie  ndnerabiUty  ^  «®  osaflator  to  symtferwbatioa  increases  with  the  i  to  C  ratio 
of  the  weiaatory  circuit.    T&m  wfe«m  tim  oeeiliators  are  to  oscillate  independently  even 
t  thwe  ia  a  «miptint;  betwwa  them,  the  L  to  C  ratk>  sbotild  be  made  as  small  as  is 
;  with    '   " "         "~~  " 


FIG.  6.    An  Oscillator  for  Low 
Harmonic  Distortion 


•  to  owHist©  at  tiie  fre^iicy  of  a  master  oscillator  without 

__ .„.._„„  „  _^  taaeter  oscffla^r,  it  is  a«f?es^ury  that  the  coupling  between 

the  aecOteum  be  uaidirwkioMl.    Tlmt  b,  the  rmetion  of  tfee  secondary  oscillator  on  the 
HM  fee  iMfliKibie.    ^^  coupling  caa  be  obtained  %  means  of  the 
CaiiUK  srwral  ^icli  amplifiers,  each  fed  by  the  master  oscillator,  the 
F  aenwmj  oscillators  ran  b@  eimtrolW  by  the  freQtiency  of  the  master  o&eilla- 

«•»  !>•  «3f»«fe3®ib«»d  at  A  subharmonie  ol  an  introduced  voltage.  The 
m$m  with  a  pifclj»roHmk  <^  the  introduced  voltages  m  no*  as  great  as 
I.  anal  ©eaaimtiowi  so  obtained  are  relatively  unstable. 


VACUUM-TUBE  OSCILLATORS  7-89 

BEAT  FREQUENCY  OSCILLATORS.  In  order  to  satisfy  the  conditions  for  stable 
oscillation  many  of  the  circuit  parameters  must  be  changed  when  the  frequency  is  changed 
over  a  relatively  large  range.  For  continuously  variable  frequency  over  a  wide  range, 
especially  over  the  audio  range,  it  is  difficult  to  make  the  necessary  changes  in  these 
parameters.  For  the  same  absolute  change  in  frequency  beginning  at  some  high  fre- 
quency, only  one  parameter,  usually  the  capacitance,  need  be  changed.  If  the  outputs  of 
two  oscillators  of  relatively  high  frequency  are  introduced  into  a  square-law  detector,  and 
if  all  the  high  frequencies  are  filtered  out,  only  the  difference  frequency  is  left.  Then  as 
the  frequency  of  one  of  these  oscillators  is  varied  from  the  frequency  of  the  other  to  that 
frequency  plus  10,000  cycles,  the  output  of  the  detector  varies  in  frequency  from  0  to 
10,000  cycles.  Such  an  arrangement  is  called  a  beat  frequency  oscillator  and  when  prop- 
erly built  is  an  excellent  laboratory  instrument. 

Care  must  be  taken  to  insure  that  the  intercoupling  between  the  oscillators  is  so  small 
that  there  is  no  tendency  for  them  to  synchronize,  since  this  will  introduce  distortion  in 
the  output  of  the  detector  due  to  the  distortion  in  the  oscillators  themselves.  This  can 
be  prevented  by  mechanical  or  electromagnetic  segregation  and  balanced  bridge  circuit 
feed  to  the  detector,  or  by  amplifiers  between  the  oscillators  and  the  detector. 

DYNATRON  OSCILLATORS.  If,  in  a  vacuum  tube,  the  grid  voltage  is  made  more 
positive  than  the  plate  voltage,  some  of  the  electrons  which  attain  a  high  velocity  between 
the  cathode  and  the  grid  pass  through  the  grid  openings,  their  velocity  carrying  them  on 
to  the  plate.  These  electrons  may  then  knock  electrons  from  the  plate.  This  process  is 
called  secondary  emission.  In  some  cases  secondary  emission  may  exist  to  such  an  extent 
that  an  increase  in  plate  voltage  actually  causes  a  decrease  in 
plate  current.  Thus,  when  a  tube  is  operated  under  these  con- 
ditions, the  a-c  plate  resistance  is  negative.  A  tube  operat- 
ing in  this  way  is  called  a  dynatron. 

An  oscillatory  circuit  connected  across  this  negative  resist- 
ance (Fig.  7)  will  oscillate  provided  the  absolute  value  of  the 

negative  resistance  is  less  than  L/rC.  Owing  to  the  small  F  "  Th  C*  *t  f 
range  of  plate  voltage  over  which  the  resistance  is  negative,  IG"  Dynatron  Oscillator0 
these  oscillators  have  not  been  made  to  give  very  large  power. 

OSCILLATORS  AT  HIGH  FREQUENCIES.  At  high  frequencies,  of  the  order  of  50 
Me,  in  the  case  of  ordinary  receiver  tubes,  where  the  time  of  transit  of  the  electrons 
between  the  tube  electrodes  becomes  an  appreciable  part  of  the  period  of  the  wave,  the 
grid  impedance  can  no  longer  be  considered  an  open  circuit  or  even  a  pure  capacitor. 
Under  these  conditions,  the  displacement  currents  arising  from  the  motion  of  the  electrons 
in  the  space  between  the  grid  and  the  other  electrodes  causes  a  true  dissipation  of  energy 
in  the  grid  circuit  within  the  tube  and  a  direct  conductive  coupling  to  other  electrodes. 
This  causes  a  grid-circuit  loading  of  the  tank  circuit,  decreasing  the  oscillator  efficiency. 
As  the  transit-time  effect  first  becomes  important,  the  resistance  of  the  grid  due  to  this 
decreases  approximately  as  the  square  of  the  frequency.  Consequently,  as  the  frequency 
of  oscillation  is  increased  through  this  frequency  range,  the  efficiency  decreases  rapidly  to 
the  point  where  oscillations  can  no  longer  be  sustained,  even  though  no  external  load  is 
coupled  to  the  tank  circuit.  Decreasing  the  dimensions  and  spacing  of  the  electrodes  in- 
creases the  frequency  at  which  transit-time  effects  become  important  but  correspondingly 
decreases  the  power-dissipating  capacity  of  the  tube. 

At  high  frequencies,  another  effect  becomes  important.  The  leads  within  the  tubes  and 
the  socket  are  found  to  have  an  appreciable  impedance.  Such  impedances  must,  of  course, 
be  considered  part  of  the  circuit  when  that  circuit  is  designed.  In  many  tube  types  the 
impedances  associated  with  the  leads  are  so  prominent  at  high  frequencies  that  the  highest 
frequency  at  which  these  tubes  will  oscillate  is  limited  by  this  consideration  rather  than 
by  the  transit-time  effect  discussed  above. 

Since  the  efficiency  of  ordinary  oscillatory  circuits  decreases  as  the  frequency  for  which 
these  circuits  are  designed  increases,  it  is  common  to  replace  coil  and  condenser  type  tank 
circuits  by  resonant  concentric  lines.  Thus  in  the  tuned-plate  tuned-grid  oscillator  circuit 
of  Fig.  l<f,  two  concentric  lines  are  used,  one  replacing  the  tuned-plate  circuit  and  one 
replacing  the  tuned-grid  circuit.  The  adjustment  of  the  resonant  frequency  is  accom- 
plished by  movable  short-circuiting  plugs  ha  the  lines.  Various  configurations  of  the  con- 
centric lines  can  be  readily  adapted  to  this  circuit. 

To  accomplish  the  combination  of  improvements  to  overcome  the  adverse  effects  of 
transit  time  and  lead  impedance,  and  to  adapt  the  tube  for  use  with  concentric  line  circuit 
elements,  the  so-called  "lighthouse"  tube  was  developed  (see  Section  4).  Using  such 
tubes,  triode  oscillators  have  been  operated  above  3000  Me. 

THE  MULTIVIBRATOR.  For  large  values  of  e  the  non-linear  theory  of  vacuum- 
tube  oscillators  discussed  above  indicates  that  voltages  having  a  discontinuous  wave  form 


7-90 


VACUUM-TUBE  CIECUIT  ELEMENTS 


-JH 


li 


resistance-capacitance  coupled  amplifiers 
connected  as  shown  in  Fig.  8,  so  that  the 
output  of  the  second  is  fed  into  the  in- 
put of  the  first.  Under  these  conditions 
the  phase  of  the  output  voltage  is  such 
as  to  aid  the  grid  of  the  input  tube.  The 
period  of  the  oscillation  is  given  roughly 
by  rgiCi  -h  J-tfCa.  Using  ordinary  tubes 
these  oscillators  can  be  made  to  operate 
at  frequencies  up  to  1,000,000  cycles. 


fm,  8.    Tls*»  UMa3  Muhfribtaior  Circuit 


BISTORTIOHLBSS  OSCILLATORS.  As  indicated  under  Non-linear  Theory  of 
OMittaton  p.  7-83,  it  is  geftfrrally  necessary  for  a  stable  oscillator  to  operate  m  such  a 
way  that  MI  increase  in  grid-volume  amplitude  would  cause  an  increase  in  limiting-type 
difltoruoa  so  that  the  voltage  developed  in  the  plate  circuit  produces  too  small  a  voltage 
is  the  grid  circuit  to  maintain  that  voltage  in  the  plate  circuit.  And  similarly  a  decrease 
in  grid-voltage  amplitude  would  produce  such  a  relatively  increased  voltage  in  the  plate 
circuit  that  a  larger  voltage  wowkJ  be  produced  in  the  grid  circuit,  tending  to  return  the 
gnd-vohage  ampKtwle  U>  its  original  value.  Under  these  conditions,  the  stable  grid- 
votage  ampittide  ia  mtfa  thai  the  voltage  developed  in  tbe  plate  circuit  is  just  sufficient 
to  maiaiA^  that  stable  grid  voltage  ia  the  grid  eircuit.  In  other  words,  the  voltage  gain 
at  tla*  oseaBbtimg  fwqtteney  must  cleerease  as  the  amplitude  increases  and  must  increase 
m  Us®  am$$£fc®&n  &&&&*%&&,  1^0  ^««*er  the  mmguitade  of  this  dependence  of  gain  upon 
amptifcacfe,  %^  «r«a^r  the  Natality  <^  tte  oscillator.  In  the  ordinary  oscillator,  where 
thie  <l&pTOd«»i»  ol  gaia  upon  amplitude  is  iMScomplislied  through  amplitude  limitmg,  the 
p^^isr  th«  dfcpefsd®&**t  tl^  greater  tli©  diBfcortkxu  Consequently,  it  is  generally  true 
that  am  foorwawi  in  atability  c^  an  o®ci!lat#r  is  accompanied  by  an  increase  in  distortion. 

IB  m&  escsEator  wter®  ^e  paa  of  the  amplifier  part  of  the  circuit  is  controlled  by  the 
mwr«®t  affitplilwie,  »fseriigt«l  ovw  one  or  more  cycles,  rather  than  by  the  relatively  in- 
•4«atAi3&©ii8  limiting  actioia  ol  ib©  ordinary  oscillator  it  is  possible  to  operate  the  amplifier 
in  tliat  part  c^  il*  Aar»«teri«ue  where  dk40rtk)i3  is  negiigiWie.  One  such  oscillator  is  the 
resa^^c^-e&pacitAziee  oscillator, 

»-C  OSCBXATOIL  Tlie  reneta«Me-eapa«itanee  tuned  oscillator  is  shown  in  Fig.  9. 
Tfee>  bride*  ««»*•  at  tlie  left  of  the  circuit  is  a  somewhat  unbalanced  Wien  bridge  at  the 
iiaeillataic  ff^faeocy,  5/a  ar v  rir^7jCj,  Tbe  wabalaace  is  such  that,  were  rj  to  be  somewhat 
,  the  bridfe  would  be  balanced.  Actually,  r0  and  r&  are  so  diosen  that  the  un- 
foh&ge  wMdb  i*  iiitrcxliieed  in  tb&  grid  carcint  of  the  first  amplifier  tube  is  just 
wfeen  ampttied  to  ppxtiKws  the  requked  bridga  suppy  voltage  between  A  and  B 
to  Bs&intAija  llutt  .^j^g^**- 

t»d>  el  tin  tsal»a«e  vt^lta^s.  i w*** j^- 

If  ti%@  gjiiii  o^  tJbe  airsplififfif 
from  its 


Tbua  such  am  osciliaior  is  amplitude  stable  at  the  frequency 
If  rjCi  »  rtf*  the  maximuni  positive  balance  voltage 
at  Mi  bwniMMBy.     Tfeer^we»  this  oscalUtor  operates  staUy  at  only  01^  fre- 
Tba  fijiia*  ex>iiut>l  b  aLverac*d  ov»?r  several  cycles  owing  to  the  thermal  lag  of 
.tAsafe  cha<R«e  <rf  HM  laxrip.    By  iiiefif is  of  g&iiged  condensers  lor  C2  and  C±1  the  f  re- 


ELECTROMECHANICAL  OSCILLATORS        7-91 

quency  of  such  an  oscillator  can  be  conveniently  varied.    Oscillators  of  this  type  can  be 
designed  for  frequencies  well  below  the  audible  limit  to  several  hundred  kilocycles. 

OSCILLATIONS  OF  GAS-FILLED  TUBES.  The  gas-filled  tube  characteristic  differs 
from  the  vacuum  tube  in  one  outstanding  particular.  The  plate  resistance  varies  from  a 
very  high  value  to  a  very  low  value  almost  discontinuously.  Such  a  characteristic  does 
not  lend  itself  to  the  production  of  sinusoidal  oscillations.  However,  it  is  highly  advan- 
tageous for  the  production  of  almost  discontinuous  wave  forms,  the  most  important  of 
which  is  the  so-called  sawtooth  wave  used  for  linear  scanning  hi  cathode-ray  oscillographs. 

The  circuit  of  such  an  oscillator  is  shown  in, Fig.  10a.  The  condenser  C  is  charged 
through  the  resistance  r  until  the  voltage  across  C  is  equal  to  the  critical  starting  voltage 
of  the  plate  circuit,  the  volt-  _ 

age  at  which  the  resistance      |  \^f\     ^        \  Rate  voltage 

suddenly  decreases  to  a  low 
value.  The  condenser  is  then 
discharged  until  the  voltage 


across  it  equals  the  critical      uH|i|l| M|l|l|l| Cb) 

stopping  voltage,  the  voltage  (aj 

at  wMch  the  plate  resistance  FIQ  ^  A  Rela^ation  0sciUator,  Using  a  Gaseous  Tube,  and  Its 
returns  to  its  high  value  Voltage  Wave  Form 

again.    This  stopping  voltage 

is  always  less  than  the  starting  voltage.  The  cycle  then  repeats  itself  as  shown  in  Fig. 
106-  These  oscillators  can  be  operated  fairly  satisfactorily  up  to  15,000  cycles,  being 
limited  by  the  ionization  and  deionization  time  of  the  gas.  Above  this  value  multivibra- 
tors must  be  used  for  linear  sweep  circuits. 

These  oscillators  are  easily  synchronized  with  an  external  frequency  by  introducing  a 
voltage  at  that  frequency  in  the  grid  circuit  as  shown  in  the  figure.  As  with  vacuum-tube 
oscillators,  these  oscillators  can  be  synchronized  at  a  subharmonic  of  the  synchronizing 
frequency. 

BARKHAUSEN  OSCILLATORS.  In  1920  Barkhausen  and  Kurz  discovered  that, 
using  certain  tubes,  oscillations  a  few  centimeters  in  wavelength  were  produced  when  the 
grid  supply  voltage  was  highly  positive  and  the  plate  was  at  or  near  the  cathode  potential 
Several  theories  of  the  operation  of  tubes  under  these  conditions  have  been  proposed  but 
none  agrees  entirely  with  the  experimental  results.  Practically  all  tubes  which  will  oscil- 
late under  these  conditions  consist  of  cylindrical  and  coaxial  cathodes,  grids,  and  plates. 
Some  investigators  have  succeeded  in  getting  oscillations  from  other  geometrical  con- 
figurations, notably  planes  instead  of  cylinders. 

Qualitatively,  the  explanation  given  is  that  electrons  rapidly  accelerated  by  means  of 
the  high  grid  potential  pass  through  the  grid  as  a  result  of  their  momentum  and  are  turned 
back  toward  the  grid,  owing  to  its  high  potential,  before  they  reach  the  plate.  This  causes 
each  electron  to  set  up  a  displacement  current  between  the  grid  and  the  plate.  A  chaotic 

distribution  of  the  phases  of  the  oscillations  of  these  elec- 
_  utpuf    £rons  can  De  shown  to  be  an  unstable  distribution.    How- 
ever, when  these  electrons  oscillate  in  clouds  the  operation 
is  stable  so  that  the  resultant  current  is  not  zero. 

The  customary  circuit  for  these  oscillators  is  shown  in 
^8*  H*    ^ke  power  of  the  oscillations  has  been  observed 
as  k*gk  as  ^  watts'  *h.e  wavelengths  go  to  6  cm,  and  the 
-D  rr        efficiencies  are  from  a  fraction  of  1  per  cent  to  7  per  cent. 

mSoi •  ^   The  lowest  efficiency  and  the  lowest  power  usually  occur  at 

the  shortest  wavelengths. 
OTHER  SPECIAL  OSCILLATORS.    Other  forms  of  oscillators  use  resonant  cavities 
for  the  oscillating  circuit  element.    These  are  treated  in  Section  4,  article  8,  on  magne- 
trons, and  article  22  of  this  section  on  cavity  resonators. 

21.  ELECTROMECHANICAL  OSCILLATORS 

The  frequency  stability  of  mechanical  vibrating  systems  is  in  general  better  than  the 
frequency  stability  of  electrical  oscillatory  circuits.  For  this  reason  mechanical  vibrating 
systems  are  coupled  to  electrical  circuits  to  give  an  electrical  output  of  constant  frequency. 

TUNING-FORK  OSCILLATORS.  The  circuit  of  a  tuning-fork  oscillator  is  shown  in 
Fig.  12.  This  type  of  oscillator  can  be  described  as  a  refined  buzzer.  The  resonant  tuning 
fork  corresponds  to  the  -oscillatory  circuit,  and  the  carbon  button  corresponds  to  the 
vacuum  tube  with  the  carbon  as  the  plate  circuit  and  the  diaphragm  as  the  grid.  The 
feedback  occurs  through  the  transformer  T.  Although  the  electric  circuit  is  tuned  by 


7-92 


VACUUM-TUBE  CIRCUIT  ELEMENTS 


O.C.  fnput 


FIG.  12.     The  Tuning  Fork  Oscillator 


means  of  the  transformer  inductances  and  the  capacitance  C,  the  harmonic  content  of  the 

output  is  high.    These  oscillators  are  very  convenient  sources  of  audio-frequency  voltage. 

MAGNETOSTRICTION  OSCILLATORS.    When  a  body  is  placed  in  a  magnetic  field, 

stresses  are  produced  within  the  body  tending  to  distort  it.    Inversely,  when  a  body  is 

distorted,  there  is  a  change  in  the  mag- 
Tuning  Fork  Carbon  Button  netic  permeability.    Magnetostriction  is 

^ ^         *      -         the  name  given  to  this  effect.     Many 

metals  and  alloys  exhibit  magnetostric- 
tion, but  according  to  Ide  it  is  most 
pronounced  in  alloys  having  8-10  per 
cent  chromium,  36-38  per  cent  nickel, 
and  the  remainder  iron  with  about  1 
per  cent  manganese  to  facilitate  forging. 
When  a  rod  of  this  material  is  mag- 
netically polarized  and  placed  in  a  coil 
carrying  alternating  current,  it  vibrates 
longitudinally  at  the  frequency  of  the 
alternating  current.  If  this  frequency  is 
the  resonant  frequency  of  the  rod  me- 
chanically, the  amplitude  of  the  effect 
will  be  large  even  for  very  small  currents 
in  the  coil.  Thus  it  can  be  considered 
equivalent  to  a  parallel  tuned  circuit  coupled  by  means  of  the  coil  into  the  electric  circuit. 
The  resonant  frequency  is  given  by  v/l,  where  v  is  the  velocity  of  sound  through  the  rod, 
about  4  km  per  sec,  and  I  is  the  length.  With  the  composition  as  above,  and  proper  heat 
treatment  and  magnetization,  the  temperature  coefficient  is  of  the  order  of  one  part  in  a 
million  per  degree  centigrade. 

These  devices  are  used  as  the  oscillatory  system  of  a  vacuum-tube  oscillator  by  means 
of  a  circuit  such  as  that  shown  in  Fig.  13.  An  additional  coil  carrying  direct  current  may 
be  necessary  if  the  plate  current  is  not  sufficient  to  polarize  the  rod.  Oscillations  are  more 
easily  controlled  when  the  relation  between  the  direction  of  the  coils  is  as  shown,  in  con- 
trast to  that  of  the  Hartley  circuit  (Fig.  la).  The  positive  feedback  is  obtained  by  means 
of  the  condenser  C  so  that  the  electrical  analogy  is  more  nearly  like  the  tuned-plate-tuned- 
grid  oscillator  shown  in  Fig.  Id.  The  rod  is  clamped  in  the  middle  (a  node  of  its  mechanical 
vibration),  and  the  ends  are  free  to  vibrate. 

The  frequency  stability  of  the  magnetostriction  oscillator  compares  favorably  with 
that  of  the  quartz  crystal  oscillators.  The  lower  limit  of  frequency  is  determined  by  the 
practical  limit  of  the  length  of  the  rod.  The  high-frequency  limit  is  due  to  the  magnetic 
skin  effect  of  the  rod.  However,  the  harmonics  of  these  high  frequencies  can  be  filtered 
out  of  the  plate  circuit  so  that  frequencies  of  several  million 
cycles  can  be  obtained  from  these  oscillators. 

PIEZOELECTRIC  CRYSTAL  OSCILLATORS.  With  respect 
to  frequency,  the  most  stable  oscillators  are  oscillators  con- 
trolled by  piezoelectric  crystals.  Crystal  oscillators  are  used 
in  practically  all  transmitters  as  master  oscillators.  A  crystal 
oscillator,  used  by  the  Bureau  of  Standards  for  the  broadcast  of 
standard  frequency  signals,  gave  a  frequency  stability  of  better 
than  1.5  parts  in  a  million  over  a  period  of  a  year,  and  its  sta- 
bility was  better  than  2  parts  in  100,000,000  over  a  period  of 
several  hours.  By  synchronizing  oscillators  to  subharmonics  of 
these  standards,  clocks  can  be  driven  which  are  considerably 

more  accurate  than  pendulum  clocks.  

Crystal  oscillators  are  applications  of  the  piezoelectric  effect,    ~     ,0    ,™    ,,        .    A  . 
which  is  a  means  of  coupling  a  mechanical  motion  to  an  electric    Fr°'     iion  (^dK 
circuit.    When  a  strain  is  produced  in  a  piezoelectric  material, 

electric  charges  appear  on  its  surface.  Conversely,  when  an  electric  field  is  produced 
between  surfaces  of  a  piezoelectric  material,  a  stress  appears  in  the  material.  Thus  if  an 
alternating  electromotive  force  is  applied  between  two  surfaces  of  the  material,  the  ma- 
terial will  vibrate,  and  if  the  material  vibrates  it  will  set  up  a  displacement  current  between 
these  surfaces. 

This  effect  has  been  observed  in  many  crystals,  the  most  important  of  which  are:  quartz, 
tourmaline,  and  Rochelle  salt  (sodium  potassium  tartrate).  Quartz  and  tourmaline  have 
been  used  in  oscillators,  quartz  crystals  dominating  the  field.  Tourmaline  is  more  expen- 
mvfc  than  qraurt*  but  has  the  advantage  that  it  can  be  ground  to  smaller  sizes  to  produce 
oscillators  having  higher  frequencies. 


ELECTEOMECHANICAL  OSCILLATORS 


7-93 


[Cgp 


For  use  in  conjunction  with  electric  circuits,  these  crystals  are  cut  in  slabs,  the  geometry 
of  which  bears  certain  relations  to  the  geometry  of  the  crystal  structure.  These  slabs  are 
mounted  in  crystal  holders,  which  consist  essentially  of  two  parallel  conducting  plates  for 
the  purpose  of  making  an  electrical  connection  with  the  surfaces  of  the  crystal.  An 
alternating  emf  set  up  between  these  plates  causes  a  current  to  flow  because  of  the  piezo- 
electric properties  of  the  crystal.  When  the  frequency  of  this  emf  is  equal  to  the  mechani- 
ical  resonant  frequency  of  the  crystal,  the  conductivity  between  the  plates  is  maximum. 
This  resonant  frequency  depends  upon  the 
dimensions  of  the  slab  and  upon  the  relation 
between  the  geometry  of  the  slab  and  the 
geometry  of  the  crystal  structure.  The  fre- 
quency of  these  resonators  is  therefore  limited 
by  the  practical  limitations  of  size  of  the  slab. 
The  frequency  limits  are  from  a  few  kilocycles 
to  a  few  megacycles.  (See  Section  13,  arti- 
cles 32-34.) 

For  the  analysis  of  electric  circuits  con- 
taining crystals  it  is  convenient  to  replace 
the  crystal  by  its  equivalent  electric  circuit. 
This  equivalent  circuit  is  simply  a  series  res- 
onant circuit  containing  resistance,  capac- 
itance, and  inductance  paralleled  by  the 
capacitance  of  the  holder.  Because  of  the 
sharpness  of  tuning  of  this  circuit,  it  is  im- 
possible actually  to  construct  the  electrical 
equivalent  circuit. 

There  are  several  oscillator  circuits  employ- 
ing crystals,  two  of  which  are  shown  in  Fig. 
14.  Circuit  (a)  makes  use  of  the  grid  to  plate 
interelectrode  capacitance  for  feedback,  as 
in  the  tuned-plate  tuned-grid  oscillator  circuit 
(Fig.  Id) ;  circuit  (6)  uses  inductive  feedback 
as  in  the  Hartley  oscillator  (Fig.  la).  As  in 


/*& 

Quartz 
Crystal 

|,|rB 

*ti 

^7 

-ilihlflu 

0 

o 

0 

^c   | 

0 

o 

0 

o 

(a)      , 

-l|l|l|l|l| 
Si 

±  !L 

FIG.  14.   Piezoelectric  Crystal  Oscillator  Circuits 


Fig.  1,  the  capacitances,  CB,  are  by-pass  condensers,  and  the  inductance,  LB,  is  a  choke 
coil  arranged  to  prevent  the  alternating  current  from  passing  through  the  d-c  supply  source, 
EB*  The  tuned  circuits  containing  L  and  C  [in  (6)  L  =  LI  +  Lz  -f  2M]  are  tuned  to 
essentially  the  resonant  frequency  of  the  crystal. 

The  grid  leaks  r#,  as  in  ordinary  vacuum-tube  oscillators,  furnish  the  operating  bias 
for  the  tubes.  The  grid  leak  is  in  this  case  limited  by  an  additional  factor,  namely,  that 
the  a-c  current  through  the  crystal,  which  is  controlled  by  this  resistance,  must  not  exceed 
the  safe  operating  value  for  the  crystal.  For  low-frequency  crystals  this  limit  is  100  ma; 
for  crystals  in  the  megacycle  range  it  reduces  to  50  ma.  Above  this  limit  the  crystal  may 
vibrate  violently  enough  to  shatter  itself.  Since  the  d-c  current  through  the  grid  leak  is  of 
the  same  order  of  magnitude  as  the  a-c  current  through  the  crystal,  an  estimate  of  the 
value  of  this  current  can  readily  be  obtained  from  the  d-c  grid  current.  In  addition  to  this 
limitation  the  correct  value  for  the  grid  leak  is  governed  by  the  operating  bias  for  the  tube 

and  varies  from  10,000  ohms  for  high-^ 
tubes  to  50,000  or  more  for  low-/x  tubes, 

The  circuit  shown  in  Fig.  14a  is  the 
most  popular  crystal-controlled  oscillator 
circuit.  Wheeler  has  analyzed  the  equiv- 
alent of  this  circuit  (Fig.  15)  by  the 
method  of  van  der  Pol  for  the  theory  of 
non-linear  oscillators.  He  obtained  certain 
criteria  for  the  frequency  to  be  dependent 

-      -An  j«    »i     primarily  upon  the  resonant  frequency  of 

nmt  Correspond^!  to    ^  ^^  ^^  ^  Q  of  the  plate  oircuit 

tuning  coil,  i.e.,  the  L  to  r  ratio,  should  be 
as  small  as  is  consistent  with  stable  oscillation.  Second,  the  plate  resistance  of  the  tube 
should  be  as  high  as  is  consistent  with  stable  oscillation.  Third,  CA,  which  includes  the 
capacitance  Cf  and  the  grid  to  filament  tube  capacitance  as  well  as  the  crystal  holder 
capacitance,  should  approach  but  not  exceed  p  —  1  times  Cgp.  Fourth,  the  plate  tuning 
capacitance,  C,  should  be  adjusted  to  give  maximum  plate  current. 

The  plate  resistance  can  be  made  large  by  the  choice  of  the  tube  and  by  decreasing  the 
operating  plate  voltage.  The  capacitance  Ck  is  usually  made  enough  smaller  than  Cgp 


te 
.  15. 


7-94 


VACUUM-TUBE  CIECUIT  ELEMENTS 


D.C.  Field  Supply 


Output 


(a  -  1)  80  that  €f  may  be  used  as  a  fine  adjustment  of  the  frequency.  The  resonant 
frequency  of  the  resonant  circuit  in  the  plate  circuit  must  be  above  the  resonant  frequency 
of  the  crystal,  while,  if  this  resonant  circuit  has  too  high  a  frequency  the  plate  load  imped- 
ance is  so  low  that  oscillations  cannot  exist.  As  the  capacitance,  C,  is  varied  from  too  low 
a  value  toward  that  value  at  which  the  resonant  frequency  of  the  plate  circuit  is  the  same 
as  that  of  the  crystal,  the  circuit  at  first  fails  to  oscillate,  then  feeble  oscillations  start 
which  increase  steadily  until  just  before  the  critical  frequency  the  amplitude  of  osculation 
rapidly  drops  to  zero.  The  total  variation  in  frequency  of  oscillation  throughout  this 
adjustment  is  2  to  5  parts  in  10,000.  The  optimum  adjustment  is  for  C  to  have  a  value 
just  less  than  that  for  critical  frequency. 

Without  some  special  means  of  temperature  control,  the  resonant  frequency  of  the 
crystal  itself  changes.  The  temperature  coefficients  of  crystals  vary,  depending  upon  the 
relation  between  the  geometry  of  the  slab  and  the  geometry  of  the  crystal  structure,  from 
1  part  in  10,000  to  1  part  in  a  million  or  less  per  degree  centigrade.  To  minimize  the 
variation  in  frequency  due  to  the  temperature,  elaborate  temperature-controlled  ovens 
are  used  to  maintain  the  temperature  constant  to  better  than  0.01  deg  cent.  For  frequency 
control  of  an  ordinary  transmitter,  ovens  capable  of  maintaining  temperatures  to  within 
0.1  deg  are  sufficiently  accurate. 

A  crystal  oscillator  may  be  designed  to  supply  as  much  as  10  watts  of  output  power  at  a 
plate  efficiency  of  30  to  60  per  cent.  The  plate  current  and  power  output  are  limited  by  the 
current  through  the  crystal.  The  usual  d-c  plate  voltage  is  from  200  to  400  volts. 

Because  of  the  low  power  output  and  in  order  to  prevent  feedback  into  the  crystal,  a 
buffer  amplifier  must  be  used  between  these  oscillators  and  the  place  where  the  power  is  to 

be  applied,  especially  when  the  load  is  variable 
as  in  modulation. 

ALTERNATORS.  Alternators  are  used  pri- 
marily to  produce  a-c  power  at  the  low  fre- 
quencies employed  in  power  engineering.  How- 
ever, before  the  advent  of  high-power  vacuum 
tubes,  comparatively  high  frequencies  for  use  in 
communication  were  obtained  from  specialized 
alternators.  Two  types  of  alternators  were 
extensively  employed  for  this  purpose. 

Alexanderson  Alternators.  The  schematic 
diagram  of  an  Alexanderson  alternator  is  shown 
in  Fig.  16.  The  toothed  wheel  is  rotated  be- 
tween the  poles  of  the  magnet  energized  by  a  d-c  field  coil.  This  changes  the  magnetic 
flux  density  through  the  output  coil  periodically,  setting  up  alternating  currents  in  the  load. 
The  frequency  of  these  alternators  is  limited  to  about  200  kc  by  the  obvious  mechanical 
limitations  of  construction  and  operation  as  well  as  by  losses  in  the  iron  core  and  teeth. 

_  Goldsdunidt  Alternators.  The  Goldschmidt  alternators  operated  on  a  different  prin- 
ciple. In  a  low-frequency  single-phase  alternator,  the  electromotive  force  developed 
in  the  armature  contains  not  only  the  fundamental  frequency  but  also  the  odd  harmonics. 
Similarly,  the  field  contains  the  even  harmonics.  Ordinarily,  the  reactances  of  the  circuits 
are  so  arranged  that  the  impedances  to  the  fundamental  and  to  the  harmonics  present  are 
relatively  high.  The  Goldschmidt  alternators  are  designed  so  that  the  impedance  at  the 
frequencies  of  these  harmonics  is  low.  Then,  by  means  of  filters,  one  of  the  higher  har- 
monics, usually  the  fourth  harmonic,  is  selected  for  use.  The  frequency  is  limited  here  to 
about  the  same  value  by  the  same  factors  as  in  the  Alexanderson  alternator 

SPARK-GAP  OSCILLATORS.  Most  of  the  early  radio  telegraph  transmitters  were 
damped-wave  transmitters.  These  damped-wave  oscillations  were  produced  by  means  of 
spark-gap  oscillators.  The  circuit  of 

one  of  these  oscillators  is  shown  in .      Spark  Gap 

Fig.  17.  When  the  instantaneous 
voltage  across  the  spark  gap  due  to 
the  audio-frequency  generator  ex- 
ceeds the  breakdown  potential  of  the 
gap,  a  sudden  rush  of  current  shock- 
exwfces  the  oscillatory  circuit  consist- 
ing of  tbe  inductance  L  and  the 
capacitance  C,  setting  up  an  oscilla- 
tory eojireot  wfeieh  is  damped  out 
bf  tbe  resistance  of  the  inductor, 


Toothed  Rotor 


FIG.  16.    Alexander-son  Alternator 


Audio 
Frequency 
Generator 


f 


Audio  Frequency 
Transformer 

FIG.  17.     Spark-gap  Oscillator 


and  by  the  load.  The  low  resistance  of  the  discharg- 
ing are  m  negligible  imtil  the  voltage  across  it  drops  below  some  comparatively  small  value 
Thw  oemri  twice  duiing  each  cycle  of  the  audio  frecpiency.  The  most  prominent  frequency 


CAVITY  RESONATOBS  7-95 

in  the  output  is  the  natural  frequency  of  the  resonant  circuit,  i.e.,  -  —  7=:*    The  out- 


put  may  be  considered  a  carrier  of  this  frequency  modulated  by  an  audio  having  a  fun- 
damental of  twice  the  frequency  of  the  generator  voltage.  There  are  so  many  harmonics 
of  the  audio  that  the  bandwidth  necessary  for  this  type  of  transmission  is  very  wide.  For 
this  reason,  the  ordinary  use  of  spark  transmitters  is  prohibited  by  law.  However,  many 
emergency  stand-by  transmitters  on  ships  are  of  this  type.  The  audio-frequency  gen- 
erator is  frequently  a  buzzer  using  the  transformer  primary  as  the  coil  so  that  these  trans- 
mitters may  be  operated  by  means  of  a  battery. 

Spark-gap  transmitters  are  actively  used  at  the  present  time  for  many  industrial  appli- 
cations, chiefly  to  supply  induction  furnaces.  The  spark  gaps  are  usually  not  ah*  gaps,  but 
a  discharge  in  mercury  vapor  alone  or  with  hydrogen.  Such  gaps  can  deliver  high  power 
at  high  efficiency,  and  they  compare  favorably  with  vacuum-tube  oscillators  for  this 
purpose. 

BIBLIOGRAPHY 

Andrew,  Victor  J.,  The  Adjustment  of  the  Multivibrator  for  Frequency  Division,  Proc.  I.R.E.,  Vol. 

19,  1911  (1934). 
Fay,  C.  E.,  and  A.  L.  Samuel,  Vacuum  Tubes  for  Generating  Frequencies  above  100  Megacycles, 

Proc.  I.R.E.,  Vol.  23,  199  (1935). 

Hayasi,  Tatuo,  The  Inner-grid  Dynatron  and  the  Duodynatron,  Proc.  I.R.E.,  Vol.  22,  751  (1934). 
Ide,  John  McDonald,  Magnetostriction  Alloys  with  Low  Temperature  Coefficients,  Proc.  I.R.E.,  Vol. 

22,  177  (1934). 
Kelster,  Frederick  A.,  Generation  and  Utilization  of  Ultra-short  Waves  in  Radio  Communication, 

Proc.  I.R.E.,  Vol.  22,  1335  (1934). 

Lapham,  E.  G.,  A  200-kilocycle  Piezo  Oscillator,  Bur.  Standards  J.  Research,  Vol.  11,  59  (1933). 
Llewellyn,  F.  B.,  Constant-frequency  Oscillators,  Proc.  I.R.E.,  Vol.  19,  2063  (1931). 
Llewellyn,  F.  B.,  Note  on  Vacuum-tube  Electronics  at  Ultra  High  Frequencies,  Proc.  I.R.E.,  Vol.  23, 

112  (1935). 
MacKinnon,  K.  A.,  Crystal  Control  Applied  to  the  Dynatron  Oscillator,  Proc.  I.R.E.,  Vol.  20,  1689 

(1932). 

Mcllwain,  Knox,  and  J.  G.  Brainerd,  High-frequency  Alternating  Currents.     John  Wiley  (1939). 
Miles,  L.  D.f  A  New  Source  of  Kilocycle  Kilowatts,  Elec.  Engg.,  Vol.  305,  54  (1935). 
Moore,  W.  H.,  Electron  Oscillations  without  Tuned  Circuits,  Proc.  I.R.E.,  Vol.  22,  1021  (1934). 
Morrison,  W.  A.,  A  High  Precision  Standard  of  Frequency,  Proc.  I.R.E.,  Vol.  17,  1103  (1929). 
Peterson,  E.,  J.  G.  Kreer,  and  L.  A.  Ware,  Regeneration  Theory  and  Experiment,  Proc.  J.B.E.,  Vol. 

22,  1191  (1934). 
Prince,  D.  C.,  and  F.  B.  Vogdes,  Vacuum  Tubes  as  Oscillation  Generators.     General  Electric  Review, 

Schenectady,  N.  Y.  (1929). 

Schneider,  E.  G.,  Radar,  Proc.  I.R.E.,  Vol.  34,  528  (1946). 
Terxnan,  F.  E.,  Radio  Engineering.  McGraw-Hill  (1932). 
Terman,  F.  E.,  R.  R.  Buss,  W.  R.  Hewlett,  and  F.  C.  Cahill,  Some  Applications  of  Negative  Feed- 

back with  Particular  Reference  to  Laboratory  Equipment,  Proc.  I.R.E.,  Vol.  10,649  (1939). 
Thompson,  B.  J.,  and  G.  M.  Rose,  Jr.,  Vacuum  Tubes  of  Extremely  Small  Dimensions  for  Use  at 

Extremely  High  Frequencies,  Proc.  I.R.E.,  Vol.  21,  1707  (1933). 
Thompson,  B.  J.,  and  P.  D.  Zottu,  An  Electron  Oscillator  with  Plane  Electrodes,  Proc.  I.R.E.,  Vol.  22, 

1374  (1934). 

van  der  Pol,  Balth,  The  Non-linear  Theory  of  Electric  Oscillations,  Proc.  I.R.E.,  Vol.  22,  1051  (1934). 
Wheeler,  L.  P.,  Analysis  of  a  Piezoelectric  Oscillator  Circuit,  Proc.  I.R.E.,  Vol.  19,  627  (1931). 

22.  CAVITY  RESONATORS 

By  L  G.  Wilson  and  J,  P.  Kinzer 

A  cavity  resonator  is  a  section  of  dielectric  completely  surrounded  by  a  metallic  shell. 
In  many  ways,  its  performance  is  analogous  to  a  resonant  R,  L,  C  low-frequency  circuit. 
However  L,  C,  and  R  can  no  longer  serve  as  basic  in  the  consideration  of  cavity  resonators, 
because  of  the  inability  to  define  inductance  and  capacitance  uniquely.  (See  reference  1.) 
In  fact  it  is  possible  to  find  only  two  such  quantities  which  describe  the  properties  of  a 
cavity  resonator. 

The  first  of  these  is  the  resonant  frequencies  (or  wavelengths)  ,  defined  as  those  values  of 
/  (or  X)  which  result  in  the  boundary  conditions  being  satisfied.  With  each  resonance 
there  is  associated  a  particular  standing-wave  pattern  of  the  electromagnetic  fields, 
These  have  been  called  "eigenvalues"  or  "eigentones,"  but  the  term  "normal  modes"  is 
now  in  general  use. 

The  second  is  the  quality  factor,  Q,  defined  by 

Energy  stored 
Energy  lost  per  cycle 


7-96 


VACUUM-TUBE  CERCTJIT  ELEMENTS 


o      o 
A      A 


o      o 
A      A 


o5         O  JH 

fill 

' 


II  ft 

.98  §.2 
n 


1 


n     n     n 


I 

n 

te! 


CAVITY  RESONATORS 


7-97 


A 
g 


o      o 
A      A 


a  s 

^  8 

g  b 

a  & 


o       o 

8      £ 


* 


2      2 


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n     n   n 

0.          qS       N 

ttj       KJ    HS 


•a    •§ 

2         2 


1 


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7-98 


VACUUM-TUBE    CIRCUIT   ELEMENTS 


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Q 

a 


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CAVITY  RESONATORS 


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or  Q  - 


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7-100 


VACUUM-TUBE    CIRCUIT   ELEMENTS 


Q 


i|j  & 
it     n     n 


-i 

I 
± 

« 

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01 

£-t 


A  H 


Approximation  for 
Tot  Number  of 
M  TE  and 
aving 


m 
u 
odes  ( 
TM)  Ha 


t 


CAVITY  RESONATORS  7-101 

In  calculating  X,  the  assumption  is  made  that  the  walls  of  the  cavity  have  perfect  con- 
ductivity. When  Q  is  calculated,  the  assumption  is  made  that  X  is  unchanged;  that  is,  the 
fields  that  actually  exist  are  those  calculated  on  the  basis  of  perfectly  conducting  walls. 
Since  the  Q's  are  generally  high,  the  approximation  is  extremely  good.  (See  reference  2.) 

Since  the  energy  is  stored  in  the  cavity  volume,  while  the  energy  is  lost  in  the  walls,  to 
obtain  a  high  Q,  the  resonator  should  have  a  large  ratio  of  volume  to  surface  area.  For 
this  reason,  cylinders,  prisms,  or  spheres  will  in  general  have  better  Q*s  than  cavities  with 
re-entrant  portions,  or  coaxial  structures. 

In  computing  the  resonant  frequencies  and  Q  values  of  cavity  resonators,  solutions  of 
the  field  equations  involve,  for  the  rectangular  prism,  circular  functions;  for  the  perfect 
cylinder,  Bessel  functions  of  the  first  kind;  and,  for  two  coaxial  cylinders,  Bessel  functions 
of  the  first  and  second  kinds.  Only  approximate  solutions  have  been  derived  for  cavities 
involving  re-entrant  sections. 

MODES.  By  fundamental  and  general  consideration,  the  modes  in  every  cavity  reso- 
nator, regardless  of  its  shape,  are  infinite  in  number  and  more  closely  spaced  as  the  fre- 
quency increases.  The  total  number,  N,  of  these  having  a  resonant  frequency  less  than 
/  is  given  approximately  by: 

N-  =  g  VJ»  (6) 

where  V  =  volume  of  cavity  in  cubic  meters. 

c  =  velocity  of  electromagnetic  waves  in  the  dielectric  in  meters  per  second. 
/  —  frequency  in  cycles  per  second. 

PRINCIPLE  OF  SIMILITUDE.  Another  theorem  generally  applicable  to  all  cavities 
is  the  principle  of  similitude,  stated  as  follows  (see  reference  3):  A  reduction  hi  all  the 
linear  dimensions  of  a  cavity  resonator  by  a  factor  1/ra  (if  accompanied  by  an  increase  in 
the  conductivity  of  the  walls  by  a  factor  m)  will  reduce  the  wavelengths  of  the  modes  by 
the  factor  \{m.  For  high-Q  cavities,  the  condition  given  in  parentheses  need  not  be 
considered. 

MODES  IN  RIGHT  CYLINDERS.  In  right  cylinders  (ends  perpendicular  to  axis) 
the  modes  fall  naturally  into  two  groups,  the  transverse  electric  (TE)  and  the  transverse 
magnetic  (TM).  In  the  TE  modes,  the  electric  lines  everywhere  lie  in  planes  perpendicu- 
lar to  the  cylinder  axis,  and  in  the  TM  modes  the  magnetic  lines  so  lie.  Further  identifi- 
cation of  a  specific  mode  is  accomplished  by  the  use  of  indices. 

THE  MS  FACTOR.  With  a  cylinder  restricted  to  a  loss-free  dielectric  and  a  non- 
magnetic surface,  the  value  of  Q  (quality  factor)  for  each  mode  depends  on  the  conduc- 
tivity of  the  metallic  surface,  the  frequency,  and  the  ratio  of  a  cross-sectional  dimension 
to  the  length.  The  quantity  Q5/X,  however,  depends  only  upon  the  mode  and  shape  and  is 
referred  to  as  the  mode-shape  (MS)  factor.  

In  this  expression,  5  refers  to  skin  depth  (see  reference  4)  in  meters  =  (l/2ir)  (Vl07p//), 
X  is  wavelength  in  meters  in  the  dielectric  ==  c//,  and  p  the  resistivity  in  ohm-meters. 
The  skin  depth  is  a  factor  which  recognizes  the  dissipation  of  energy  in  the  walls  and  ends 
of  the  cylinder.  With  increase  of  resistivity  of  these  surfaces  the  currents  penetrate 
deeper  and  the  resulting  Q  is  lower. 

FUNDAMENTAL  FORMULAS.  Expressions  for  standing-wave  patterns  and  Qd/\ 
are  given  in  Table  1,  for  right  rectangular,  circular,  and  full  coaxial  cylinders.  The  mode 
indices  are  Z,  m,  n  following  the  notation  of  Barrow  and  Mieher  (reference  5).  In  the 
rectangular  prism  they  denote  the  number  of  half-wavelengths  along  the  coordinate  axes. 
For  the  other  two  cylinders  they  have  an  analogous  physical  significance  with  Z  related  to 
the  angular  coordinate,  m  to  the  radial,  and  n  to  the  axial. 

In  an  elliptical  cylinder,  a  further  index  is  needed  to  distinguish  between  modes  which 
differ  only  in  their  orientation  with  respect  to  the  major  and  minor  axes;  these  paired 
modes  are  termed  even  and  odd  and  have  slightly  different  resonant  frequencies  (see 
reference  6) .  In  the  circular  cylinder  they  have  the  same  frequency,  a  condition  which  is 
referred  to  as  a  degeneracy  (in  this  case,  double) ;  that  is,  in  the  circular  cylinder,  odd  and 
even  modes  are  distinguishable  only  by  a  difference  in  their  orientation  within  the  cylinder 
with  reference  to  the  origin  of  the  angular  coordinate.  In  Table  1,  the  field  expressions 
are  given  for  the  even  modes;  those  for  the  odd  modes  are  obtained  by  changing  cos  Id  to 
sin  W  and  sin  IB  to  cos  IB  everywhere. 

The  value  of  N  in  the  table  for  the  circular  cylinder  is  based  on  counting  this  degeneracy 
as  a  single  mode;  counting  even  and  odd  modes  as  distinct  will  nearly  double  the  value  of 
N,  thus  bringing  it  into  agreement  with  the  general  eq.  (6) . 

In  Table  1,  the  mks  system  of  units  is  implied.  The  notation  is  in  general  accordance 
with  that  used  in  prior  developments  of  the  subject.  For  engineering  applications,  it  is 
advantageous  to  reduce  the  results  to  units  in  ordinary  use  and  to  change  the  notation 


7-102 


VACUUM-TUBE  CIRCUIT  ELEMENTS 


wherever  this  leads  to  a  more  obvious  association  of  ideas.  For  these  reasons,  in  what 
follows  attention  is  confined  to  the  right  circular  cylinder,  with  changes  in  units  and 
notation  as  specified  later. 


4,0 


FIG.  18.     Mode  Chart  for  Circular  Cylinder  Cavity  Resonator.    Z>  and  L  in  inches-  /  in  megacycles 
per  second.    The  rectangle  indicates  the  best  operating  region  for  the  TE  Oil  mode, 

THE  MODE  CHART.    The  formula  relating  the  resonant  frequency  to  the  mode, 
shape,  and  dimensions  may  be  written  simply: 

C/D}*  =  A  +  Bn*  (2V  (7) 

frequency  in  megacycles  per  second. 
D  =»  diameter  of  cavity  in  inches. 
L  =  length  of  cavity  in  inches. 
4  *  a  csoastaat  depending  upon  the  mode.    Values  of  A  are  given  in  Table  2  for 

Ifee  kwesfe  30  modes.    Values  of  Bessel  function  roots  are  given  in  Table  3 

lor  fafc  180  modes. 


CAVITY  RESONATORS 


7-103 


B  =  a  constant  depending  upon  the  velocity  of  electromagnetic  waves  in  the  dielec- 
tric. For  air  at  25  deg  cent  and  60  per  cent  relative  humidity,  B  =  0.34799  X 
108. 

n  -  third  index  defining  the  mode,  i.e.,  the  number  of  half  wavelengths  along  the 
cylinder  axis. 

Formula  (7)  represents  a  family  of  straight  lines,  when  (D/£)2  and  (/D)2  are  used  as  co- 
ordinates, and  leads  directly  to  the  easily  constructed  and  highly  useful  mode  chart  of 
Fig.  18. 

It  will  be  noted  from  Table  2  that  the  TE  Qmn  and  the  TM  Imn  modes  have  the  same 
frequency  of  resonance.  This  is  a  highly  impor- 
tant case  of  degeneracy.  In  the  design  of  prac- 
tical cavities  it  is  necessary  to  take  measures  to 
eliminate  this  degeneracy,  as  the  TM  mode  (usu- 
ally referred  to  as  the  companion  of  its  associated 
TE  mode)  introduces  undesirable  effects. 

DESIGN  OF  HIGH-Q  CAVITY  IN  TE  Oln 
MODE.  In  many  applications,  a  resonator  is 
used  in  its  fundamental  (gravest)  mode.  How- 
ever, when  high  values  of  Q  are  desired,  it  may 
be  necessary  to  use  a  high-order  mode.  In  this 
case,  it  is  desirable  to  keep  the  volume  the  mini- 
mum because  other  modes  can  cause  undesired 
responses  and  other  deleterious  effects.  As  shown 
by  eq.  (6),  the  total  number  of  resonances  is  a 
function  of  the  volume.  Analysis  of  the  problem 
leads  to  the  conclusion  that  operation  in  the  TE 
Oln  mode  (unimportant  exceptions  occur  for 
values  of  Q5/\  less  than  1.2)  gives  the  smallest 
volume  for  an  assigned  Q  and  also  leads  to  specific 
values  of  n  and  D/L  which  give  this  result.  In 
fact,  for  maximum  Q  per  volume  in  the  TE  Oln 
mode, 


Table  2.     Constants  for  Use  in  Com- 
puting   the    Resonant   Frequencies    of 
Circular  Cylinders 


B 


•  0.34799  X  10s 
1.17981  X  1010  in.  per  sec 


=  3.11  X  108 


(8) 


which  permits  easy  plotting  on  a  mode  chart  of 
the  locus  of  the  operating  points  for  best  Q  per 
volume  ratio. 

The  mode-shape  (MS}  factor  for  the  TE  Oln 
modes  may  be  expressed  as  follows: 


2.77 


1  +  0.168     — 


This  has  been  derived  from  Table  1  by  com- 
bining terms  which  are  a  function  of  frequency 
and  by  assuming  the  conductivity  of  copper 
(p  =  1.7241  X  10~8  -  the  International  Stand- 
ard value  for  copper)  for  the  cylinder  walls. 

The  relative  Q's  for  several  metals  are:  silver 
1.03,  copper  1.00,  gold  0.84,  aluminum  0.78,  and 
brass  0.48.  Therefore,  a  brass  cavity  will  have 


Mode 

r 

A 

TM  01 

2.40483 

0.  81563  X  10s 

02 

5.52008 

4.2975 

03 

8.65373 

10.5617 

n 

3.83171 

2.0707 

12 

7.01559 

6.9415 

13 

10.17347 

14.5970 

21 

5.13562 

3.7197 

22 

8.41724 

9.9923 

31 

6.38016 

5.7410 

32 

9.76102 

13.4374 

41 

7.58834 

8.1212 

51 

8.77148 

10.8511 

61 

9.93611 

13.9238 

TE  01 

3.83171 

2.0707 

02 

7.01559 

6.9415 

03 

10.17347 

14.5970 

11 

1.84118 

0.47810 

12 

5.33144 

4.0088 

13 

8.53632 

10.2770 

21 

3.05424 

1.3156 

22 

6.70613 

6.3426 

23 

9.96947 

14.0175 

31 

4.20119 

2.4893 

32 

8.01524 

9.0606 

41 

5.31755 

3.9879 

42 

9.28240 

12.1520 

51 

6.41562 

5.8050 

61 

7.50127 

7.9359 

71 

8.57784 

10.3772 

81 

9.64742 

13.1265 

Value  of  c  is  for  air  at  25  deg  cent  and  60 
per  cent  relative  humidity.  D  and  L  in 
inches ;  /  in  megacycles. 


about  one-half  of  the  Q  of  a  similar  copper  cavity.  Silver-plating  a  copper  cavity  will 
increase  Q  about  3  per  cent.  Experience  shows  that  only  80  to  90  per  cent  of  the  theo- 
retical Q  can  be  realized.  This  should  be  taken  account  of  in  the  design. 

With  the  frequency  and  desired  Q  known,  the  dimensions  of  the  cavity  can  be  deter- 
mined. 

CAVITY  COUPLINGS.  To  be  useful  the  cavity  must  be  coupled  to  external  circuits. 
The  coupling  to  all  modes  can  be  analyzed,  at  least  qualitatively,  from  the  field  expres- 
sions of  Table  1.  The  problem  is  to  get  the  correct  coupling  to  the  desired  mode  and  as 
little  coupling  as  possible  to  all  others.  This  may  be  obtained  either  by  a  loop  or  a  probe 
at  the  end  of  a  coaxial  line  or  by  an  orifice  connecting  the  cavity  with  a  wave  guide.  For 
optimum  coupling  the  plane  of  a  loop  must  be  perpendicular  to  the  H  lines;  the  axis  of  a 


7-104 


VACUUM-TUBE  CIRCUIT  ELEMENTS 


probe  must  be  colHnear  with  the  E  lines;  and  the  H  lines  in  a  wave  guide  feeding  through 
an  orifice  must  be  parallel  to  the  H  lines  in  the  cavity.  ^ 

Since  the  electric  field  is  zero  everywhere  at  the  boundary  surface  of  the  cavity  for  the 
TE  Oln  mode,  coupling  to  it  must  be  magnetic;  a  probe  cannot  be  used.    The  location  for 

Table  3.    Values  of  the  Bessel  Function  Zero  (rZm)  for  the  First  180  Modes  in  a  Circular 

Cylinder  Resonator 


Tim 

Mode* 

rim 

Mode 

rjm 

Mode 

rjm 

Mode 

\     1.8412 

E      1-1 

46  13,0152 

M     3-3 

91   18.4335 

M  10-2 

136  22.6716 

E      2-7 

2     2.4048 

M     0-1 

47  13.1704 

E      2-4 

92  18.6374 

E      6-4 

f!37  22.7601 

M     1-7 

5     3.0542 

E     2-1 

M8  13.3237 

M     1-4 

93  18.7451 

E    12-2 

I  138  22.7601 

E      0-7 

(    4     3.8317 

M     1-1 

I  49   13.3237 

E     0-4 

94  18.9000 

M  14-1 

139  22.9452 

M     8-4 

{    5     3.8317 

E      0-1 

50  13.3543 

M     9-1 

95  18.9801 

M     5-4 

140  23.1158 

M  14-2 

6     4  .  20  1  2 

E      3-1 

51    13.5893 

M     6-2 

96  19.0046 

E      9-3 

141   23.2548 

E    21-1 

7     5.1356 

M     2-1 

52  13.8788 

E    12-1 

97  19.1045 

E    17-1 

142  23.2568 

M  18-1 

8     5.3176 

E      4-1 

53  13.9872 

S      5-3 

98  19.1960 

E     4-5 

143  23.2643 

E    16-2 

9     5.3314 

E      1-2 

54  14.1155 

S      8-2 

99   19.4094 

M    3-5 

144  23.2681 

E      7-5 

10     5.5201 

M    0-2 

55  14.3725 

M    4-3 

100  19.5129 

E      2-6 

145  23.2759 

M  H-3 

11     6.3802 

M     3-1 

56  14.4755 

M  10-1 

101   19.5545 

M     8-3 

146  23.5861 

M     6-5 

12     6.4156 

E      5-1 

57  14.5858 

E      3-4 

f  102  19.6159 

M     1-6 

147  23.7607 

E    10-4 

13     6.7061 

E      2-2 

58  14.7960 

M    2-4 

1  103  19.6159 

E      0-6 

148  23.8036 

E      5-6 

f  14     7.0156 

M     1-2 

59  14.8213 

M     7-2 

104  19.6160 

M  11-2 

149  23.8194 

E    13-3 

(  15     7.0156 

E     0-2 

60  14.8636 

E      1-5 

105  19.8832 

E    13-2 

150  24.0190 

M     4-6 

16     7.5013 

E      6-1 

61    14.9284 

E    13-1 

106  19.9419 

E      7-4 

151   24.1449 

E      3-7 

17     7.5883 

M     4-1 

62  14.9309 

M     0-5 

107  19.9944 

M  15-1 

152  24.2339 

M     9-4 

18    8.0152 

S      3-2 

63  15.2682 

S      6-3 

108  20.1441 

E    18-1 

153  24.2692 

M  15-2 

19     8.4172 

U     2-2 

64  15.2867 

B      9-2 

109  20.2230 

E    10-3 

154  24.2701 

M     2-7 

20     8.5363 

E      1-3 

65  15.5898 

M  11-1 

110  20.3208 

M     6-4 

155  24.2894 

E    22-1 

21     8.5778 

S      7-1 

66  15.7002 

M     5-3 

111  20.5755 

E      5-5 

156  24.3113 

E      1-8 

22     8.6537 

M    0-3 

67   15.9641 

E      4-4 

112  20.7899 

M  12-2 

157  24.3382 

M  19-1 

23     8.7715 

M    5-1 

68  15.9754 

E    14-1 

113  20.8070 

M     9-3 

158  24.3525 

M     0-8 

24     9.2824 

E      4-2 

69  16.0378 

M     8-2 

114  20.8269 

M    4-5 

159  24.3819 

E    17-2 

25     9.6474 

E      8-1 

70  16.2235 

M     3-4 

115  20.9725 

E      3-6 

160  24.4949 

M   12-3 

26     9.7610 

M     3-2 

71   16.3475 

E      2-5 

116  21.0154 

E    14-2 

161   24.5872 

E      8-5 

27     9.9361 

M     6-1 

72  16.4479 

E    10-2 

117  21.0851 

M  16-1 

162  24.9349 

M     7-5 

28     9.9695 

IS      2-3 

f  73  16.4706 

M     1-5 

118  21.1170 

M     2-6 

163  25.0020 

E    14-3 

{  29  10.1735 

M     1-3 

174  16.4706 

E      0-5 

119  21.1644 

E      1-7 

164  25.0085 

E    11-4 

}  30  10.1735 

E      0-3 

75  16.5294 

E      7-3 

120  21.1823 

E    19-1 

165  25.1839 

E      6-6 

31    10.5199 

E      5-2 

76  16.6982 

3f  12-1 

121  21.2116 

Jf    0-7 

166  25.3229 

J£    23-1 

32  10.7114 

E      9-1 

77  17.0038 

M     6-3 

122  21.2291 

j£      8-4 

167  25.4170 

M  16-2 

33  11.0647 

M     4-2 

78  17.0203 

S    15-1 

123  21.4309 

E    11-3 

168  25.4171 

M  20-1 

34  11.0864 

M     7-1 

79  17.2412 

Jtf     9-2 

124  21.6415 

M     7-4 

169  25.4303 

M     5-6 

35  11.3459 

£      3-3 

80  17.3128 

E      5-4 

125  21.9317 

E     6-5 

170  25.4956 

E    18-2 

36  11.6198 

M     2-3 

81   17.6003 

#    11-2 

126  21.9562 

M   13-2 

171   25.5094 

Jf   10-4 

37  11.7060 

^      1-4 

82  17.6160 

M     4-4 

127  22.0470 

If  10-3 

172  25.5898 

#      4-7 

38  11.7349 

S      6-2 

83  17.7740 

J£      8-3 

128  22.1422 

E    15-2 

173  25.7051 

M  13-3 

39  11.7709 

E    10-1 

84  17.7887 

.E      3-5 

129  22.1725 

M  17-1 

174  25.7482 

Jf     3-7 

40  11.7915 

Jfcf     0-4 

85  17.8014 

M  13-1 

130  22.2178 

M     5-5 

175  25.8260 

E      2-8 

41    I2.225T 

M     8-1 

86  17.9598 

jfcf     2-5 

131  22.2191 

#    20-1 

176  25.8912 

J£      9-5 

42  12.3386 

If     5-2 

87  18.0155 

E      1-6 

132  22.4010 

tf      4-6 

f  177  25.9037 

Af     1-8 

43  12.6819 

E      4-3 

88  18.0633 

£    16-1 

133  22.5014 

E      9-4 

I  178  25.9037 

E      0-8 

44  12.8265 

s  u-r 

89  18.0711 

Jfef     0-6 

134  22.5827 

Jf     3-6 

179  26.1778 

J£    15-3 

45  12.9324 

£      7-2 

90  18.2876 

M     7-3 

135  22.6293 

E    12-3 

180  26.2460 

E    12-4 

*  Nomenclature  after  Barrow  and  Mieher,  Natural  Oscillations  of  Electrical  Cavity  Resonators 
Free.  I.R.E.,  April  1940,  p.  184. 

M  modes  take  zeros  of  Jj<x) ;  ^  modes  take  zeros  of  Jz'fcc).  Number  directly  following  E  or  AT  is  Z; 
number  after  hyphen  is  number  of  root. 

Values  less  than  16.0  are  abridged  from  six-place  values  and  are  believed  to  be  correct;  values  more 
than  16.0  are  abridged  from  five-place  values  and  may  be  in  error  by  one  unit  in  fourth  decimal  place. 
Underlined  5  in  fourth  place  indicates  that  higher  value  is  to  be  used  in  rounding  off  to  fewer  decimals." 

maximum  coupling  is  on  the  side  of  the  cavity,  an  odd  number  of  quarter-guide  wave- 
lengths from  the  end,  or  on  the  end  about  halfway  (48  per  cent)  out  from  the  center  to  the 
ed«©.  Correct  orientation  requires  the  axis  of  a  loop  to  be  parallel  to  the  axis  of  the 
cylmder  for  side-wail  feed  and  to  be  perpendicular  to  the  cylinder  axis  for  end  feed.  Wave- 
guide orientation  is  shown  in  Table  4. 

Tfee  theory  of  coupling  loops  and  orifices  is  not  at  present  precise  enough  to  yield  more 
tfean  approximate  c&rmnsions.  On  the  basis  of  rather  severely  limiting  assumptions, 
coupling  fonaiiias  for  a  round  hoie  connecting  a  rectangular  wave  guide  and  a  TE  Oln 


CAVITY  RESONATORS 


7-105 


CD 

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CD 
cu 


QOH13W  QNHdnOO 


UJ  UJ 

2  2 


I 


Q 
u 


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—     CO    O 

s  2  a 


c  o    -  o 

fc   •*    rt  o> 

£    w    —  q 

d   d  d 

So  N 

53  P 

£  S  S  2 

odd 


^ » s  a 

O  —  <vi 

i  §  § 

UJ  UJ  UJ 

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U.  ?  O 


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7-106  VACUUM-TUBE   CIRCUIT  ELEMENTS 

cavity  are  given  on  Table  4.  The  assumptions  are  that  the  orifice  is  in  a  wall  of  negligible 
thickness,  its  diameter  is  small  compared  to  the  wavelength,  it  is  not  near  any  surface 
discontinuity,  and  the  wave  guide  propagates  only  its  principal  (gravest)  mode  and  is 
perfectly  terminated.  In  some  applications,  the  computed  diameter  is  somewhat  smaller 
than  experiment  shows  to  be  correct. 

Coupling  by  means  of  an  electron  beam  can  also  be  used,  but,  owing  to  transit  time,  a 
re-entrant-type  cavity  is  usually  used  to  keep  down  the  distance  the  electrons  must  travel 
within  the  resonator. 

BIBLIOGRAPHY 

1.  Hanaen,  W.  W.,  and  R.  D.  Richtmyer,  On  Resonators  Suitable  for  Klystron  Oscillators,  J.  Applied 

2.  HanaSn,'  W?  W.%  Type  of  Electrical  Resonator,  J.  Applied  Phys.,  Vol.  9,  654-663  (1938).         . 

3.  K6nig,  H.f  The  Laws  of  Similitude  of  the  Electromagnetic  Field,  and  Their  Application  to  Cavity 

Resonators,  Hochf.  tech  u.  Elek:akust  Vol.  58,  174-180  (1941).     Also  Wireless  Engineer,  Vol.  19, 
216-217,  No.  1304  (1942). 

4.  Wheeler,  H.  A.,  Formulas  for  the  Skin  Effect,  Proc.  I.R.E.,  Vol.  30,  412-424  (1942). 

5.  Barrow,  W.  L.,  and  W.  W.  Mieher,  Natural  Oscillations  of  Electrical  Cavity  Resonators,  Proc. 

6.  Chu,  L'j,,  Electromagnetic  Waves  in  Elliptical  Hollow  Pipes  of  Metal,  /.  Applied  Phys.,  9,  583- 

591  ( 193S) 

7.  Borgnis,  F.,  Elektromagnetisehe  Eigenschwingungen  dielektriseher  Raume,  Ann.  Physik,  Vol.  35, 

35&-3S4  (1939). 

a  Borgnis,  F.t  Die  konzentrische  Leitung  als  Resonator,  Hoctif.  tech  u.  Elek:akus,  Vol.  56,  47-54 
(1940).  Resonant  modes  and  Q  of  the  full  coaxial  resonator.  For  long  abstract,  see  Wireless 
Engineer,  Vol.  18,  23-25  (1941). 

9.  Wilson,  I.  G.,  G.  W.  Schramm,  and  J.  P.  Kinzer,  High  Q  Resonant  Cavities  far  Microwave  Test- 
ing. B.S.T.J.,  Vol.  25,  408-434  (1946). 

10.  Kinzer,  J.  P.,  and  I.  G.  Wilson,  End  Plate  and  Side  Wall  Currents  in  Circular  Cylinder  Cavity 

Resonator.    B.S.T.J.  Vol.  26,  31-79  (1947). 

11.  Kinzer,  J.  P.,  and  I.  G.  Wilson,  Some  Results  on  Cylindrical  Cavity  Resonators.    B.S.T.J.  Vol. 

26,  410-445  (1947). 

POWER  SUPPLY 

By  J.  E.  Young 

23.  RECEIVER  POWER  SUPPLY 

The  first  receivers  to  employ  vacuum  tubes  used  batteries  for  filament  and  plate  power 
supplies,  but  tubes  were  soon  developed  which  had  cathodes  suitable  for  heating  by 
alternating  current.  At  the  same  time  rectifier-filter  combinations  were  developed  to  sup- 
ply plate  voltage,  and  it  was  no  longer  necessary  to  depend  on  batteries,  where  a-c  power 
was  available.  During  the  ensuing  years  the  size  of  batteries  has  been  reduced,  and  their 
shape  has  been  adapted  to  receiver  applications.  At  the  same  time  their  life  has  been 
considerably  increased.  Batteries  are  now  used  extensively  in  receivers  in  areas  where  no 
power  is  available  and  for  portable  receivers  which  are  often  designed  so  that  either  bat- 
tery or  a-c  power  may  be  used.  Sealed  storage  batteries  have  also  been  developed,  and 
receivers  using  them  arranged  so  that,  if  the  receiver  is  plugged  in  an  a-c  line,  the  batteries 
are  automatically  recharged. 

FILAMENT  POWER.  -  The  filament  power  for  a-c  operated  receivers  is  usually  ob- 
tained from  one  or  more  windings  of  the  receiver  power  transformer.  The  design  of  power 
transformers  is  covered  on  pp.  $-26  to  6-30.  A  common  winding  is  customarily  employed 
to  excite  all  cathodes;  however,  it  sometimes  happens  that  both  filamentary  and  indirectly 
heated  cathode  tubes  are  used  in  a  receiver,  and  bias  is  applied  to  the  filamentary  tubes 
in  such  a  way  that  the  filament  winding  of  the  transformer  is  maintained  at  a  potential 
considerably  above  ground.  It  is  possible,  in  this  case,  that  the  difference  in  potential 
between  the  cathode  and  heater  of  the  indirectly  heated  tubes  will  exceed  the  value 
specified  by  the  tube  designer.  Where  this  condition  exists,  or  where,  because  of  the 
circuits  in  which  the  cathodes  of  the  tubes  are  connected,  it  may  likewise  be  possible  to 
exceed  the  rated  potential  difference  between  the  cathode  and  the  heater  for  which  the 
tube  was  designed,  it  is  necessary  to  provide  separate  filament  windings,  grouping  the 
tubes  on  these  windings  in  such  a  manner  that  excessive  voltage  strain  will  not  exist 
between  cathode  and  heater  of  any  tube. 

MATE  POWER.  The  voltage  supply  for  the  plate  circuits  of  the  receiving  tubes  in 
a  modem  receiver  is  usually  obtained  from  a  vacuum-tube  rectifier.  A  high  a-c  voltage 
is  applied  to  the  tube,  and  the  rectified  output  is  passed  through  a  low-pass  filter  to  atten- 
uate the  a~c  components.  In  Fig.  1  a  typical  B  supply  circuit  is  shown.  With  a  condenser 


RECEIVER  POWER  SUPPLY 


7-107 


f 

Fl 

3O  f 

Ctic 

lenry 

"kes 

•nfflRHT* 

Z2          5 

i 

J~ 

C3 

A*. 

\ 
\ 
\ 

I 

Filament 
<r  —  r!>-T- 
Suppfy 

t 
Voltage 

B     1               i        o 

T4=  1   = 

UJ 

•)oiooJ 

j.    l—  » 

!        \ 


115  Volts 
(Rms.) 

Rectifier 


122  Vi'V  I 
Tap        | 


iFor  Choke  Inpat 
Omit  Ct 


•  Fitter  - 


-Qatptrt  Systei 


FIG.  1.     Typical  Receiver  Power  Supply 


60^Applied  Voltage 


Current  In 
1st  Plate 


Current  in 
2nd  Plate 


5afhode 
Current 


60/v.ApplLed  Voliage 


Cuaenf  in  1st  Plate 


Current  In  2nd  Plate 


Cathode  Cm-cent 


Condenser  Input  Choke 

FIG.  2.     Wave  Form  in  Rectifier  Circuits 


7-108 


VACUUM-TUBE   CIRCUIT  ELEMENTS 


input  to  the  B  filter  (Ci  connected)  the  ratio  of  d-c  output  voltage  to  a-c  input  voltage  is 
higher  than  with  a  choke  input  (Ci  disconnected).  The  choke  input  has  the  advantage  of 
lower  peak  currents  with  less  danger  of  damage  to  the  tube  under  overload  conditions. 
The  voltage  regulation  with  variable  load  is  also  better  with  choke  input. 

In  Fig.  2  the  form  of  the  currents  and  voltages  is  given  for  various  portions  of  these 
two  circuits. 

Figures  ZA  and  SB  illustrate  the  relations  between  output  voltage  and  load  current  for 
a  typical  full-wave  rectifier  for  a  choke  input  filter  circuit  and  for  a  condenser  input  filter 


circuit. 


520 


40  80          120 

D,C.  Load  MHfiamperes 

A.  Choke  Input  to  Filter. 


12.0 


40.          80          120        160 
D.C-  Load  Mliriamperes 

B.  Condenser  Input  to  Filter. 


FIG.  3.    Output  Characteristics  of  Rectifier  Tubes 

MERCURY- VAPOR  RECTIFIERS.  The  efficiency  of  rectification  is  improved  by  the 
presence  of  mercury  vapor  in  the  tube.  The  mercury  vapor  reduces  the  voltage  drop  in 
the  tube  and  improves  the  voltage  regulation.  This  type  of  tube  may  produce  r-f  inter- 
ference unless  special  precautions  are  taken,  such  as  shielding  of  the  rectifier  tube  and  the 
use  of  r-f  chokes  or  resistors  hi  the  rectifier  plate  leads. 

B  SUPPLY  FILTER.  The  capacitor  input  filter  is  the  most  economical  type  if  the 
transformer  impedance  and  rectifier  tube  type  are  such  that  excessive  charging  current 
will  not  be  experienced.  In  general,  electrolytic  capacitors  are  compact  and  inexpensive 
enough  so  that  filters  are  usually  designed  with  a  rather  large  ratio  of  capacitance  to  in- 
ductance. It  frequently  happens  that  tubes,  or  the  elements  of  tubes  used  for  different 
purposes  in  the  receiver,  require  different  voltages.  In  this  event  it  is  economical  to  ob- 
tain additional  filtering  by  using  the  voltage  drop  resistors  and  shunt  capacitors  as  a 
resistance-capacitor  filter.  Since  the  capacitive  reactance  will  generally  be  small  com- 
pared to  the  resistance,  the  attenuation  of  this  type  of  filter  is  inversely  proportional  to 
the  product  of  the  resistance  and  capacitance.  If  this  is  so,  sufficient  filtering  is  provided 
in  the  main  filter  to  produce  the  required  ripple  attenuation  for  the  tubes  which  are  sup- 
plied directly  from  this  filter. 

It  is  possible  to  obtain  increased  attenuation  by  substituting  a  shunt  resonant  circuit 
for  the  inductive  element  of  the  filter;  however,  this  practice  is  rarely  followed,  since  the 
attenuation  of  ripple  frequencies  higher  than  the  resonant  frequency  of  the  circuit  is 
reduced  and  it  is  costly,  in  production,  to  maintain  the  values  of  inductance  and  capaci- 
tance closely  enough  to  insure  resonance  at  the  correct  frequency.  The  speaker  field  is 
frequently  used  as  one  of  the  chokes  of  the  filter,  and  occasionally  a  hum-bucking  coil  is 
provided  on  the  voice  coil  of  the  loudspeaker. 


24.  TRANSMITTER  POWER  SUPPLY 

A-C  POWER.  A-c  power  is  used  for  filament  and  plate  supplies  for  transmitters 
almost  universally.  Filaments  are  usually  heated  by  alternating  current  directly ;  rectifier- 
filter  systems  are  provided  for  plate  and  bias  supplies.  Where  alternating  current  is  not 
available,  motor  generators  or,  for  very  low  power  transmitters,  batteries  are  used. 


TRANSMITTER  POWER  SUPPLY  7-109 

FILAMENT  POWER.  Small  transmitting  tubes,  unless  used  in  service  requiring 
quick-heating  types,  usually  have  indirectly  heated  cathodes.  Unipotential  cathodes 
are  also  required  to  permit  operation  at  very  high  radio  frequencies.  Considerations 
affecting  the  design  of  filament  power  supplies  for  such  tubes  are  covered  in  Section  6,  ar- 
ticle 14.  Large  tubes  are  usually  directly  heated  and  have  either  a  tungsten  filament  or, 
where  practicable,  a  thoria-coated  tungsten  filament.  The  emission  efficiency  of  the  latter 
is  higher  and  less  filament  power  is,  therefore,  required  to  achieve  a  given  space  current. 

Filament  power  is  usually  derived  from  a  115-  or  230-volt  bus  connected  to  the  trans- 
mitter supply  through  a  regulator,  rheostat,  or  other  means  of  holding  the  filament  voltage 
within  required  limits  (generally  ±5  per  cent),  and  necessary  control  switches  and  protec- 
tive devices.  Step-down  transformers  convert  the  bus  voltage  to  a  value  suitable  for 
application  to  the  filaments  of  the  tubes.  It  should  be  noted  that  the  life  of  tungsten- 
filament  tubes  is  greatly  affected  by  small  changes  in  voltage.  It  is  advisable,  therefore, 
to  provide  means  of  adjusting  the  filament  voltage  of  each  such  tube  independently.  The 
filament  voltage  should  be  no  higher  than  is  required  to  provide  the  necessary  emission 
current. 

FILAMENT  STARTING.  The  cold  resistance  of  transmitting-tube  filaments  is  gen- 
erally less  than  a  tenth  of  the  hot  resistance.  Filament-heating  currents  are  usually  large 
enough  to  cause  severe  mechanical  stresses  in  the  filaments  and  their  supports.  It  is 
necessary,  therefore,  to  limit  the  filament  starting  current  to  a  safe  value.  This  is  de- 
termined by  the  designer  of  the  tube  and  is  frequently  specified  as  one  and  one-half  times 
the  normal  running  current.  Starting-current  limitation  is  secured  either  by  applying  the 
filament  voltages  in  steps,  controlled  so  that  the  current  reaches  a  steady  value  before 
the  next  voltage  increment  is  applied,  or  by  using  a  current-Umiting  reactor  in  the  filament 
transformer  primary  circuit.  The  necessary  reactance  may  be  designed  into  the  filament 
transformer  itself,  or  a  separate  reactor  may  be  used.  This  method  of-  limiting  filament 
current  is  preferable  since  the  voltage  increase  across  the  filament  terminals  is  smooth  and 
is  directly  controlled  by  the  filament  resistance. 

HUM  DUE  TO  FILAMENT  CURRENT.  The  electron  stream  emitted  by  the  fila- 
ment is  affected  by  the  magnetic  field  set  up  around  the  filament  by  the  heating  current. 
This  effect  is  a  periodic  change  in  the  space  impedance  between  the  filament  and  each  of 
the  other  elements  of  the  tube,  at  twice  the  frequency  of  the  filament  exciting  current. 
The  magnitude  of  this  effect  is  a  function  of  the  design  of  the  tube.  The  transmitter 
designer  may  minimize  it  either  by  the  use  of  a  tube  having  a  multistrand  filament  con- 
nected to  a  two-  or  three-phase  heating  source,  or  by  heating  the  filaments  of  tubes  con- 
nected in  pushpull  or  parallel  from  different  phases  of  a  two-  or  three-phase  power  supply. 
Tubes  connected  in  pushpull  and  delivering  power  to  the  load  at  different  parts  of  the 
audio-frequency  cycle  should  have  their  filaments  heated  by  currents  which'  are  in  phase. 
The  reduction  of  ripple  obtained  by  multiphase  filament  connection  will  generally  be  of 
the  order  of  10  db  for  a  two-phase  filament  connection  and  14  db  to  18  db  for  three-phase. 
The  improvement  obtained  by  the  use  of  more  than  three  phases  is  small,  and  the  in- 
creased sensitivity  of  the  ear  to  the  higher  resulting  ripple  frequency  may  make  the  hum 
more  objectionable. 

PLATE  POWER.  Except  for  emergency  or  mobile  equipment,  a-c  power  supplies  are 
generally  used.  The  a-c  potential  is  rectified  and  filtered  to  obtain  direct  current  having 
the  requisite  freedom  from  ripple. 

TYPES  OF  RECTIFIERS.  Selenium  or  copper  oxide  rectifiers  are  frequently  used  to 
obtain  the  relatively  low  voltages  required  for  bias  or  for  the  plate  circuits  of  low  power 
tubes.  For  high-voltage  plate  supplies,  rectifier  tubes  are  employed.  These  may  be  of 
several  types.  The  most  common  are  vacuum  tubes  having  a  hot  cathode  emitter  and  a 
cold  plate.  To  neutralize  the  space-charge  drop,  a  small  quantity  of  mercury  or  one  of 
the  inert  gases  is  introduced  into  the  tube.  Other  types  of  rectifier  tubes  use  a  pool  of 
mercury  as  the  cathode.  Electron  emission  is  obtained,  either  by  maintaining  an  electron 
discharge  from  a  hot  spot  on  the  surface  of  the  pool  by  means  of  an  auxiliary  electrode 
excited  by  a  separate  power  source  or  by  discharging  a  heavy  current  through  a  concen- 
trated point  on  the  surface  of  the  mercury  by  means  of  an  electrode  which  just  touches 
the  surface  of  the  mercury.  The  first  of  these  two  types  of  tubes  has  been  called  the  pool- 
type,  mercury-arc  rectifier;  the  second  is  known  as  the  Ignitron.  The  pool-type  tube  is 
available  for  single-  or  multiphase  operation,  whereas  the  Ignitron  is  commonly  a  single 
rectifier  unit  and  a  number  are  connected  in  groups  for  multiphase  operation.  The  steel- 
tank  mercury-arc  rectifier  has  been  frequently  used  for  high-power  applications  such  as 
railway  power  supplies.  It  has  also  been  used  to  a  considerable  extent  in  Europe  as  a 
source  of  high  voltage  for  radio  transmitters.  Because  of  its  relatively  high  cost  and  the 
necessity  for  the  provision  of  considerable  auxiliary  equipment,  it  has  not  been  widely 
used  in  radio  transmitters  in  the  United  States. 


7-110  VACUUM-TUBE   CIECUIT  ELEMENTS 

25.  RECTIFIER  CIRCUITS 

Figure  4  shows  the  common  rectifier  circuits  for  Kenotron  and  mercury-vapor  tube 
type  rectifiers  with  their  voltage  and  current  conditions.  In  calculating  the  d-c  output 
voltage,  from  the  "E  average"  given  for  the  circuits  shown,  the  voltage  drop  across  the 
filter  choke  and  the  tube  drop  should  be  subtracted  from  E  average  to  arrive  at  the  d-c 
output  voltage.  The  relations  hold  if  a  filter  choke  large  enough  to  insure  a  constant  load 
current  is  assumed.  In  practice,  such  a  choke  is  usually  required  to  obtain  adequate  ripple 
attenuation.  Eectifier  tubes  and  steel-tank  rectifiers  are  rated  on  the  basis  of  the  max- 
imum average  plate  current  they  can  carry, 'the  duration  of  the  cycle  over  which  the  plate 
current  flows,  and  the  total  voltage  to  which  the  tube  is  subjected  during  the  period  when 
it  is  not  passing  current.  This  voltage  is  known  as  the  peak  inverse  voltage.  In  the  hot- 
cathode,  mercury-vapor  tube,  the  safe  peak  plate  current  is  also  given,  since,  owing  to  the 
low  internal  tuba  drop,  dangerously  high  currents  may  be  passed  through  the  tube,  result- 
ing in  rapid  deterioration  of  the  filaments. 

DOUBLE  OUTPUT  RECTIFIERS.  In  many  transmitter  applications  it  is  desirable 
to  provide  two  plate  voltages,  one  for  the  low  power  stages  and  the  other  for  the  output 
amplifiers.  If  a  single  rectifier  is  to  be  used,  the  low  voltages  may  be  obtained  by  means  of 
dropping  resistors.  However,  if  they  are  equal  to  or  less  than  one-half  the  rectifier  output 
voltage,  one  of  the  rectifier  circuits  which  permits  obtaining  half- voltage  is  more  economi- 
cal. The  single-phase  bridge  circuit  or  the  three-phase  full-wave  circuit  shown  in  Fig.  4, 
columns  2  and  5  respectively,  may  be  so  used.  Half-voltage  is  obtained  from  a  center  tap 
on  the  secondary  of  the  plate  transformer  in  the  single-phase  full-wave  circuit,  or  from 
the  center  of  the  wye-connected  secondaries  in  tjie  three-phase  full-wave  circuit.  The 
additional  transformer  kHovolt-amperes  required  for  the  half-voltage  loads  can  be  com- 
puted by  considering  the  equivalent  rectifier  circuits,  which  are  the  center-tapped  circuit 
shown  in  column  1  for  the  single-phase  rectifier  or  the  three-phase  half-wave  circuit  shown 
in  column  3  for  the  three-phase  rectifier.  If  the  amplitude  of  the  current  to  be  supplied 
at  half-voltage  is  an  appreciable  part  of  the  total,  a  d-c  component  of  current  is  produced 
in  the  secondaries  of  the  high-voltage  transformers,  in  the  three-phase  rectifier,  which 
may  seriously  affect  their  operation.  The  effect  of  the  d-c  component  may  be  eliminated 
by  using  a  two-winding  secondary  for  each  phase  of  the  wye,  connected  in  broken  star. 
It  is  generally  desirable  to  use  separate  niters  for  the  two  voltage  outputs  in  this  type  of 
rectifier;  otherwise  objectionable  interaction  may  result,  causing  higher  ripple  voltage 
output  than  anticipated,  or  low-frequency  feedback  oscillation. 

HIGH-VOLTAGE  TRANSFORMERS.  Transformers  used  in  rectifier  service  are  spe- 
cially designed  (see  Section  6,  article  14),  since  the  insulation  requirements  and  heating 
effects  are  quite  different  from  those  experienced  in  a-c  circuit  practice.  Secondary  wind- 
ing insulation  will  depend  on  the  type  of  rectifier  circuit  used.  In  the  single-phase  center- 
tapped  secondary  circuit  shown  in  column  1,  for  instance,  the  center  tap  of  the  secondary 
winding  is  usually  at  substantially  ground  potential,  whereas  in  the  single-phase  full-wave 
circuit  shown  in  column  2  the  midpoint  of  the  secondary  winding  is  at  a  potential  equal 
to  half  the  d-c  voltage  developed  by  the  rectifier.  Fault  conditions  will  also  seriously 
affect  the  insulation  requirement  of  the  transformer.  In  any  of  the  circuits  in  which  the 
center  tap  or  midpoint  of  the  wye  of  the  transformer  secondaries  is  connected  to  ground 
through  the  filter  choke,  a  short-circuit  fault  will  momentarily  cause  the  full  d-c  voltage 
to  be  developed  across  the  filter  choke,  raising  the  potential  of  the  center  point  of  the 
transformer  secondaries  to  the  full  d-c  voltage  above  ground. 

Transformers  for  three-phase  rectifier  circuits  may  be  constructed  as  a  single  three- 
phase  unit,  or  three  separate  single-phase  transformers  may  be  used.  The  first  cost  of 
the  unit  transformer  will  generally  be  lower,  but  having  three  single-phase  transformers 
permits  temporary  operation  in  an  open  delta  circuit  if  one  transformer  fails.  Trans- 
formers may  be  obtained  for  rectifiers  rated  up  to  several  hundred  kilowatts  and  up  to 
10,000  to  15,000  volts  d-c  output,  in  either  the  dry  or  oil-filled  types.  Oil-filled  trans- 
formers for  any  power  are  available,  filled  with  Transil  oil  or  one  of  the  non-inflammable 
oils  sold  trader  various  trade  names,  such  as  Pyranol  or  Dykanol.  The  non-inflammable 
oils  require  special  transformer  designs,  since  they  will  attack  some  of  the  insulation 
materials  ordinarily  used.  In  installations  subject  to  the  rules  of  the  insurance  under- 
wifcers,  it  is  usually  necessary  that  transformers  filled  with  inflammable  insulating  oil 
be  Jammted  in  separate  fireproof  vaults  provided  with  oil  sumps  and  drains.  The  size 
<*f  the  transformer  for  which  such  protection  must  be  provided  is  determined  by  its  oil 
eontenfc  sad  varies  in  the  different  states.  Local  codes  should  be  checked. 

FILTER  DESIGN.  The  design  of  the  filter  depends  largely  on  the  service  for  which 
tin©  transmitter  is  to  be  used.  Telegraph-transmitter  filters  are  designed  to  the  require- 


RECTIFIER   CIRCUITS 


7-111 


-H 


o 


O 

eo 


COO3 

H 
+  1 


p 
'6 


9 


COCO 
CO  CD 

+ 1 


O     W  |M 


7-112  VACUUM-TUBE  CIRCUIT  ELEMENTS 

ment  that  the  load  current  may  vary  at  a  rate  corresponding  to  the  telegraph  characters, 
while  the  filters  for  telephone  transmitters  must,  usually,  be  capable  of  supplying  power 
at  a  very  low  audio  frequency.  It  is  usually  necessary  first  to  design  the  filter  to  secure 
the  desired  ratio  between  the  load  voltage  and  ripple  voltage  and  then  determine  whether 
it  fulfils  other  requirements. 

Let  F  be  the  principal  ripple  frequency.  Then  F  =  supply  frequency  times  number  of 
rectifier  phases.  Let  xc  =  filter  capacitive  reactance;  XL  =  filter  inductive  reactance; 
C  —  filter  capacitance  in  microfarads,  and  L  »  filter  inductance  in  henrys. 

Single  stage: 

Per  cent  ripple  = (1) 

XL  —  xc 

Double  stage: 

Per  cent  ripple  =  7 r«  (2) 

(XL  —  xcr 

where  xc  and  XL  are  the  capacitive  and  inductive  reactances,  respectively,  of  each  of  two 
similar  stages;  m  =  70  for  single-phase  full-wave  rectifier,  24  for  a  three-phase  half-wave 
rectifier,  and  5  for  a  three-phase  full-wave  rectifier. 

Most  Economical  Filter  Design.  A  single-stage  filter  is  more  economical  than  the 
double-stage  type  unless  an  unusually  large  reduction  in  ripple  is  desired,  or  the  frequency 
is  low,  or  low  filter  choke  reactance  is  necessary,  as  for  a  telegraph  transmitter. 

Inductance-capacitance  Ratio.  The  ratio  between  filter  inductance  and  capacitance 
depends  on  a  number  of  factors.  If  no  other  requirements  are  imposed  on  the  filter  than 
ripple  reduction,  the  most  economical  ratio  of  L  to  C  may  readily  be  calculated.  Ordi- 
narily, however,  the  LC  ratio  is  fixed  by  other  considerations.  If  the  rectifier  tubes  are 
worked  near  their  current  rating,  the  extra  current  flowing  through  the  tube  due  to  the 
impedance  of  the  filter  should  be  checked  to  determine  whether  the  total  tube  current  is 
excessive.  This  component  of  tube  current  may  be  calculated  as  follows : 

rms  ripple  voltage 

r      ...   f^ 

XL  —  xc 

Filter  for  Telephone  Transmitter.  In  most  telephone  transmitters  the  filter  must 
supply  an  audio  component  of  power,  since  the  time  lag  through  the  filter  and  transformer 
reactances  is  too  great  to  permit  the  audio  component  to  be  drawn  directly  from  the  trans- 
former. The  frequency  and  amplitude  of  the  audio  component  depend  on  the  type  of 
modulation.  The  relations  for  the  various  systems  are  given  below: 

1.  Class  B  Audio:  Rectifier  is  required  to  supply  an  audio  component  having  a  peak 
value  equal  to  the  difference  between  the  no-signal  and  maximum  instantaneous  signal 
plate  currents  of  the  class  B  stage  at  a  rate  corresponding  to  twice  the  lowest  transmitted 
audio  frequency. 

2.  Linear  Amplifier  and  Grid-bias-modulated  Amplifier:  Rectifier  is  required  to  supply 
an  audio  component  having  a  peak  value  equal  to  the  unmodulated  plate  current  multi- 
plied by  the  modulation  factor. 

3.  Constant-current:  No  audio  component  exists  in  the  d-c  power  source. 

It  will  be  seen  from  the  above  considerations  that  the  linear  amplifier  and  the  grid-bias- 
modulated  amplifier  impose  the  severest  restrictions  on  filter  design,  while  the  constant- 
current  system  requires  the  filter  to  supply  no  audio-frequency  power.  The  class  B 
modulator  requires  some  audio  power,  but  the  facts  that  the  lowest  audio  frequency 
existing  in  the  filter  is  twice  the  lowest  modulating  frequency  and,  further,  that  the  class  C 
modulated  amplifier  is  usually  supplied  from  the  same  source,  drawing  a  steady  current  in- 
dependent of  the  modulation  frequency,  make  the  filter  design  somewhat  easier  in  this 
case. 

Linear  Amplifier.  If  we  assume  a  linear  amplifier,  completely  modulated,  and  it  is 
desired  to  find  a  suitable  filter  combination,  the  following  method  may  be  used 
(if  r*  >  L/C}:  Let  F0  =  resonant  frequency  of  L  and  C;  Fd  »  frequency  at  which* distor- 
tion begins;  F  =  any  audio  frequency,  to  be  investigated;  K  «  ratio  of  load  voltage  at 
peak  of  audio  cycle  to  unmodulated  load  voltage;  K$  =  above  ratio  at  frequency  F<z;  r  = 
load  resistance;  L  =  filter  inductance;  and  C  =  filter  capacitance.  Then 

K  =  IT^fT  (4) 


RECTIFIER  CIRCUITS  7-113 

L  ^  W  - 
r         ZvFFfK 


Fd  «  FQVKd  +  1  (6) 

K*  =  1  -  ^  (7) 

Sample  Calculation.  Assume  that  a  linear  amplifier  requires  an  output  of  10  kw  at  15,000  volts- 
A  filter  consisting  of  a  15-henry  inductance  and  7.5-pf  capacitor  is  provided.  Determine  the  resonant 
frequency  of  the  filter,  the  value  of  K  at  30  cycles,  the  frequency  at  which  distortion  begins,  and  K  at 
that  frequency. 


=  15  cycles  per  second 


L  =  15  henrys 

C  -  7.5  id  =  7.5  X  10~6  farad 


6.28\/15  X  7.5  X  10~6 
F  =  30  cycles  per  second 
a  =  jp02  -  F2  «  225  -  900  -  -675 

47r2F2Fo*L2  =  4  X  9.85  X  900  X  50,600  X  225 
r2  506,000,000 

K  =  ~675  -  -0.999 

V465.000  +  797 

^  =  i-4- 


506.000,000  X  7.5  X  10 


Fd  =  15V0.99605  +1  =21.2  cycles  per  second 

Thus,  it  will  be  seen  that,  at  30  cycles,  the  reduction  in  rectifier  voltage  at  the  peak  of  the  audio  cycle 
is  only  0.1  per  cent,  and  distortion  due  to  the  filter  circuit  will  not  be  encountered  at  any  frequency 
above  21.2  cycles.  At  this  frequency  the  reduction  in  voltage  is  0.395  per  cent. 

Class  B  Modulator.  The  same  method  of  calculation  may  be  followed  for  a  class  B 
modulator,  except  that  some  allowance  should  be  made  for  the  steady  d-c  component  of 
plate  current  supplying  the  no-signal  plate  current  for  the  modulator  and  the  modulated 
r-f  amplifier.  An  approximate  method  is  to  find  K  as  above  and  then  to  find  the  actual 
ratio,  use  the  equation  K'  =  2  —  /(I  —  K}  —  K,  where  Kf  is  the  ratio  of  load  voltage  at 
the  crest  of  an  audio  cycle  to  the  unmodulated  load  voltage,  corrected  for  the  unmodu- 
lated component  of  plate  current  representing  the  sum  of  the  class  C  amplifier  plate  cur- 
rent and  the  no-signal  plate  current  of  the  modulator;  /  is  the  ratio  of  peak  load  current 
to  load  current  without  modulation;  and  K  is  calculated  in  the  same  manner  as  for  a 
linear  amplifier.  It  must  also  be  remembered  that  only  the  double  frequency  component 
of  the  modulating  frequency  appears  in  the  rectifier  circuit,  and  so  the  modulating 
frequency  should  be  doubled  when  it  is  used  to  evaluate  FQ,  F<f,  and  F. 

Filter  Chokes.  Methods  of  calculation  of  the  inductance  required  for  the  filter  choke 
are  cove-red  in  the  section  on  filters.  In  addition  to  inductance,  it  is  necessary  to  specify  the 
voltage  insulation  or  type  of  construction,  the  d-c  current,  and  the  a-c  voltage.  In  normal 
operation  the  full  ripple  voltage  of  the  rectifier  will  appear  across  the  terminals  of  the 
choke.  Its  winding  insulation  must,  therefore,  be  sufficient  to  withstand  this  voltage. 
In  addition,  a  short  circuit  in  the  load  will  subject  the  choke  to  the  full  d-c  rectifier  output 
voltage,  so  that  it  must  be  designed  to  withstand  this  strain.  The  voltage  insulation  re- 
quired between  winding  and  core  for  normal  operation  will  be  lowest  if  the  choke  is  con- 
nected between  the  rectifier  and  ground.  This  does  not,  of  course,  eliminate  the  necessity 
for  the  provision  of  adequate  insulation  to  take  care  of  load  short-circuit. 

Rectifier-tube  Operation.  The  hot-cathode  mercury-vapor  tube  is  used  in  the  majority 
of  high-voltage  rectifiers  for  radio  transmitters.  In  addition  to  the  limits  of  peak  inverse 
voltage,  and  peak  and  average  current  set  up  in  the  rating  of  each  of  these  types  of  tubes, 
it  is  also  necessary  that  the  condensed  mercury  temperature  be  maintained  within  specified 
limits.  For  operation  of  the  tubes  at  their  maximum  rated  peak  inverse  voltage,  it  is 
usually  necessary  to  keep  the  condensed  mercury  temperature  between  the  limits  of  20 
and  60  deg  cent.  For  temperature  ranges  extending  from  10  to  70  deg  cent,  the  maximum 
peak  inverse  voltage  is  frequently  halved.  To  control  the  condensed  mercury  tempera- 


7-114  VACUUM-TUBE  CIRCUIT  ELEMENTS 

ture  a  jet  of  air  is  directed  against  a  spot  on  the  lower  edge  of  the  glass  bulb,  and  the 
temperature  of  this  air  stream  is  occasionally  controlled  by  means  of  auxiliary  heaters. 
It  is  sometimes  desirable  to  secure  additional  current  capacity  by  connecting  rectifier 
tubes  in  parallel.  Because  of  the  peculiar  conduction  characteristics  of  gases,  unless  such 
tubes  are  identical  the  one  in  which  the  gas  is  ionized  first  will  conduct  all  the  current  and 
the  other  will  not  break  down.  This  condition  may  be  corrected  by  connecting  a  small 
center-tapped  choke  between  the  two  tubes  or  by  connecting  a  resistance  in  series  with 
each  tube  so  that  sufficient  potential  is  available  to  break  down  the  second  tube  after  the 
first  one  has  started  conducting. 

TUBE  HEATER  DELAY.  Most  of  the  hot-cathode  mercury-vapor  tubes  use  highly 
efficient  shielded  cathods  or  filaments  in  order  to  reduce  the  filament  power  to  a  minimum. 
Such  cathodes  require  some  time  to  come  up  to  their  operating  temperature,  and  it  is 
usually  advisable  to  provide  a  time-delay  relay  to  prevent  the  accidental  application  of 
plate  voltage  before  the  cathode  has  reached  its  operating  temperature.  For  the  same 
reason,  and  to  prevent  the  adherence  of  any  particles  of  mercury  to  the  anode  or  cathode, 
each  new  tube  should  be  baked  out  thoroughly  before  plate  potential  is  applied,  and  then 
reduced  plate  voltage  should  be  applied,  slowly  working  up  to  the  normal  operating  plate 
voltage.  Unless  these  precautions  are  followed,  severe  arc-backs  may  result  and  the  tube 
will  be  permanently  damaged. 

TUBE-FAILXTRB  PREDICTION.  Mercury-vapor  rectifier  tubes  almost  always  fail 
by  arcing  back,  that  is,  becoming  conductive  to  a  voltage  of  either  sign.  This  condition 
will  occur,  momentarily,  and  then  clear  itself;  however,  as  the  tube  ages,  it  happens  with 
increasing  frequency,  until  it  cannot  be  tolerated,  and  the  tube  must  be  replaced.  Each 
arc-back  short-circuits  the  plate  transformer  and  usually  trips  the  a-c  overcurrent  relays. 
If  a  bank  of  tubes  is  used  in  a  multiphase  rectifier  it  is  difficult  to  determine  by  visual 
observation  which  tube  has  arced  back.  Devices  which  will  register  the  flow  of  reverse 
current,  such  as  polarized  magnetic  drops,  are  sometimes  used  as  indicators.  However, 
the  short-circuit  current  is  often  so  great  that  it  will  cause  the  indicators  to  drop  on  other 
tubes  as  well  as  on  the  defective  one.  It  is  possible  to  predict  with  fair  accuracy  the  time 
when  a  tube  may  be  expected  to  fail  by  making  a  routine  check  of  the  arc-drop  voltage 
when  the  tube  is  carrying  rated  current.  When,  on  successive  readings,  separated  by 
perhaps  100  hours  of  normal  operation,  the  arc  drop  is  found  to  be  rising  rapidly,  the 
tube  will  probably  soon  fail  and  should  be  removed  from  service.  These  tests  must,  of 
course,  be  made  by  removing  the  tube  from  its  operating  position,  and  applying  the  necessary 
test  voltage,  which  need  not  be  greater  than  100  wits. 

RECTIFIER  CONTROL  SYSTEMS.  Since  high-power  rectifiers  must  usually  be  de- 
signed to  have  low  regulation,  a  fault,  in  the  form  of  either  a  short  circuit  in  the  load  or  an 
arc-back  in  a  rectifier  tube  or  tank,  may  result  in  dangerously  high  currents  in  the  system. 
To  minimize  any  trouble  resulting  from  this  source,  a  high-speed  breaker  should  be  pro- 
vided in  the  power  transformer  primary.  The  breaker  should  be  controlled  by  a-c  over- 
load relays  in  each  phase  of  the  primary  and  by  a  d-c  overload  relay  in  the  output  circuit. 
If  a  short  circuit  should  occur  in  the  transmitter,  the  energy  stored  in  the  filter  will  be 
dissipated  in  the  fault  even  though  the  primary  circuit  is  cleared  instantly.  For  this 
reason  it  is  advisable  to  incorporate  some  series  resistance  in  the  load  circuit  to  aid  in 
dissipating  the  filter  energy.  A  resistor  of  1  to  5  per  cent  of  the  load  resistance  can  usually 
be  added  with  no  bad  effects.  It  should  be  remembered,  in  designing  such  a  resistor,  that 
for  a  gassy  tube  or  similar  fault  the  load  resistance  is  virtually  zero,  and  all  the  rectifier 
voltage  will,  for  an  instant  at  least,  appear  across  the  protective  resistor.  This  should 
h'ave  sufficient  thermal  capacity  to  dissipate  several  times  the  energy  stored  in  the  filter 
and  should  be  insulated  to  cany  the  full  rectifier  voltage  across  its  terminals.  As  a  further 
protection  against  high  voltages  across  the  power  transformer  secondary  in  the  event  of 
an  arc-back,  it  is  advisable  to  connect  a  spark  gap  in  series  with  a  current-limiting  resistor 
between  each  high-potential  secondary  terminal  and  ground.  The  gap  may  take  the  form 
of  either  a  horn  or  sphere  gap  and  should  be  set  to  break  down  at  about  1.5  times  the 
normal  voltage. 

BIBLIOGRAPHY 

Armstrong,  R.  W.,  Polyphase  Rectification  Special  Connections,  Proc.  I.R.B.,  January  1931. 
Lee,  R.,  Rectifier  Filter  Circuits,  Elec&ic  J.,  Vol.  29,  April,  1932. 
Lee,  R,,  Radio  Telegraph  Keying  Transients,  Proc.  IM,&,  Vol.  22  (February  1934). 
Prmce  and  Vogdes,  If ercury  Arc  Rectifiers  and  Circuits.    McGraw-Hill  (1927). 


RADIO   RECEIVERS  7-115 

RADIO  RECEIVERS 

By  Vernon  D.  Landon 

The  functions  of  a  radio  receiver  are  to : 

First:  Select  a  desired  signal  from  the  heterogeneous  signals  picked  up  by  the  antenna. 

Second:  Amplify  the  radio-frequency  signal  selected. 

Third:  Detect  the  signal,  thereby  producing  audio-frequency  currents.  (In  the  case  of 
continuous  wave  code  signals,  it  is  necessary  to  heterodyne  the  signal  with  a  local  oscillator 
before  detecting.) 

Fourth:  Amplify  the  audio-frequency  signal. 

Fifth:  Reproduce  the  signal  audibly  by  means  of  a  loud  speaker  or  headphones. 

The  parts  of  a  receiver  performing  the  above  functions  sometimes  have  overlapping 
duties.  For  example,  the  antenna  circuit  gives  some  amplification  due  to  resonance  and 
has  some  selectivity. 

The  simplest  antenna  coupling  circuit  is  shown  in  Fig.  1JL,  with  its  equivalent.  ra 
and  Ca  are  the  effective  resistance  and  capacitance  of  the  antenna,  and  Ls  and  rg  are  the 
inductance  and  resistance,  of  a  variable  inductor,  in  the  receiver.  Ea  is  the  voltage  in- 
duced in  the  antenna  by  the  incoming  signal.  The  step-up  ratio  of  the  circuit  is  denned 
as  the  ratio  of  E8  to  Ea.  At  resonance  (neglecting  that  component  of  Es  due  to  r»,  E8/Ea 
—  juLs/r,  where  r  =  r0  +  r«.  The  step-up  at  a  frequency  other  than  resonance  is  E's/Era 
=  /wZ/8/z,  where  z  —  ra  +  rs  -\-  j(uLs  —  1/coC). 

The  ratio  of  the  step-up  at  resonance  to  that  at  a  frequency  differing  from  resonance 
by  a  given  amount  is  known  as  the  selectance,  or  the  discrimination  ratio,  for  the  given 
frequency  difference.  A  curve  of  selectance  vs.  frequency  difference  is  a  selectivity  curve. 

To  a  rather  close  approximation  the  selectance  is  equal  to  S  =  I  -f-  j4.TrfdL/r,  where  fd 
measures  the  frequency  difference  from  resonance.  Since  the  only  circuit  constants  in 
this  expression  are  L  and  r,  the  figure  L/r  is  said  to  determine  the  selectivity  of  the  circuit. 
The  selectivity  is  not  changed  by  a  change  of  carrier  frequency  if  L/r  is  kept  constant. 

The  circuit  of  Fig.  \A  has  several  disadvantages.  The  step-up  is  high  and  reasonably 
constant  over  the  tuning  range,  but  the  selectivity  is  poor,  owing  to  the  large  antenna 
resistance  in  series  with  the  tuned  circuit.  Also  it  is  very  difficult  to  incorporate  in  a 
unicontrol  tuning  system.  The  circuits  of  Figs.  1JE?,  1C,  and  ID  are  more  commonly  used. 

In  \B  a  tunable  circuit  is  connected  to  the  antenna  through  a  small  coupling  condenser 
Cc.  If  Cc  is  quite  small  (as  it  is  in  practice)  then  the  antenna  resistance  and  capacitance 
may  be  neglected  with  only  a  slight  error.  By  the  use  of  Thevenin's  theorem  the  circuit 
then  reduces  to  that  on  the  right  of  Fig.  IB.  This  is  a  simple  series  circuit.  The  step-up 
of  such  a  series  circuit,  considered  by  itself,  is  nearly  a  constant  over  the  tuning  range. 
It  is  exactly  constant  if  rs  is  exactly  proportional  to  the  frequency.  In  practice  rs  usually 
varies  slightly  more  rapidly  than  the  frequency.  However,  in  this  case  the  input  voltage 
varies  with  frequency.  As  indicated  in  the  diagram  the  effective  driving  voltage  E'a 
=  EaCc/(C3  +  Cc).  Since  Cs  +  Cc  is  the  capacitance  which  produces  resonance,  then 
C4  -f-  Cc  is  inversely  proportional  to  the  square  of  the  frequency.  Hence  E'a.  is  directly 
proportional  (and  the  output  voltage  Es  is  also  roughly  proportional)  to  the  square  of  the 
frequency.  This  is  the  chief  disadvantage  of  this  circuit.  Its  advantages  are  its  good 
selectivity  and  the  ease  with  which  it  may  be  incorporated  in  a  unicontrol  tuning  system. 

In  Fig.  1C  the  tunable  circuit  is  connected  to  the  antenna  through  a  large  inductance. 
If  this  coupling  inductance  were  quite  large  the  capacitance  and  resistance  of  the  antenna 
could  again  be  neglected.  If  the  power  factors  of  Le  and  -Ls  are  assumed  to  be  equal  a 
transformation  involving  Th£veninjs  theorem  gives  the  equivalent  circuit  shown.  Here, 
since  Ls  and  Le  are  constant,  the  input  voltage  Ea  is  constant.  If  all  these  assumptions 
are  correct  and  rs  is  proportional  to  the  frequency,  the  output  voltage  E8  is  also  constant. 
In  practice  there  are  three  effects  combining  to  produce  a  marked  drooping  of  the  step-up 
at  the  high-frequency  end  of  the  tuning  range.  First,  the  inductance  of  Lc  is  usually  not 
large  enough  to  make  the  antenna  capacitance  negligible.  Hence,  the  effective  inductance 
of  Lc  and  the  antenna  in  series  is  less  at  low  frequencies.  This  increases  the  low-frequency 
step-up.  Second,  the  distributed  capacitance  of  Le  increases  its  effective  inductance 
most  at  high  frequencies  and  lowers  the  high-frequency  step-up.  Third,  the  resistance  of 
T8  usually  varies  faster  than  the  first  power  of  the  frequency,  lowering  the  high-frequency 
step-up.  . 

An  antenna  circuit  which  is  often  used  involves  a  combination  of  capacitative  and  in- 
ductive coupling  in  order  to  obtain  a  flat  step-up  characteristic.  This  circuit  is  shown  in 
Fig.  ID,  with  its  equivalent.  It  may  consist  of  a  tuned  secondary,  of  the  usual  type, 
coupled  to  a  primary  of  about  eight  times  the  secondary  inductance.  Loose  inductive 


7-116  VACUUM-TUBE  CIRCUIT  ELEMENTS 


i 

Es 

T 


i 

E* 

J 


A 


o 


O 


O. 

-D- 


FIG,  1.    Antenna  Coupling  Circuits 


TYPES  OF  RECEIVERS 


7-117 


coupling  is  used.  The  capacitative  coupling  is  adjusted  to  the  value  required  to  give  the 
desired  step-up  at  the  high-frequency  end  of  the  tuning  range.  For  uniform  and  max- 
imum step-up  it  is  essential  that  the  inductive  coupling  have  the  proper  phase,  so  that 
the  capacitative  coupling  adds  rather  than  subtracts.  To  obtain  this  condition  the 
grid  and  antenna  leads  must  emerge  from  the  transformer  with  opposite  directions  of 
rotation.  The  step-up  of  such  a  transformer  is  practically  constant  over  its  tuning  range. 
In  a  transformer  for  the  broadcast  band  the  value  of  the  step-up  usually  lies  between  the 
limits  of  3  and  10,  depending  on  design  constants.  If  tight  coupling  is  used  giving  high 
step-up,  the  penalty  is  more  detuning  with  changes  in  antenna  constants.  This  results  in 
poor  tracking,  with  the  other  tuned  circuits  of  the  receiver,  unless  an  antenna  is  used  of 
the  size  for  which  the  circuit  was  designed. 


26.  TYPES  OF  RECEIVERS 

CRYSTAL  DETECTOR  RECEIVER.  The  simplest  type  of  complete  receiver  is  a 
crystal  detector  circuit  such  as  shown  in  Fig.  2.  The  selectivity  of  this  receiver  is  very 
slight,  and  amplification  is  lacking  except  for  that  due  to  resonance.  Nevertheless  local 
stations  can  be  received. 

For  low-impedance  crystals,  the  selectivity  of  the  receiver  can  be  improved  by  con- 
necting the  input  to  the  crystal  across  a  portion  of  the  coil. 


FIG.  2.     Crystal  Detector  Receiver    » 


FIG.  3.    Regenerative  Receiver 


REGENERATIVE  RECEIVER.  Figure  3  shows  the  circuit  of  a  regenerative  detector. 
The  inductive  coupling  of  the  small  coil  in  the  plate  circuit  to  the  tuning  inductance  is 
adjustable.  When  the  regenerative  feedback  is  adjusted  to  a  critical  value,  just  less  than 
that  required  to  produce  self-oscillation,  a  great  amplification  of  signals  results.  The 
selectivity  curve  is  much  too  sharp,  resulting  in  a  loss  of  sidebands,  thus  unduly  impairing 
the  fidelity  of  reproduction.  The  greatest  objection  to  regenerative  receivers  is  due  to 
their  ability  to  oscillate  when  the  feedback  is  too  great.  This  results  in  a  radiated  signal 
which  produces  very  objectionable  squeals,  or  beat  notes,  in  near-by  receivers.  This  type 
of  receiver  is  now  illegal.  It  can  be  made  legal  by  placing  a  neutralized  stage  of  r-f  amplifi- 
cation ahead,  so  that  it  will  not  radiate  when  properly  shielded. 

Superregeneration  of  the  Blocking  Type.  If  the  feedback  in  Fig.  3  is  advanced  well 
beyond  the  point  of  oscillation,  the  oscillations  become  self-modulated.  This  is  due  to  a 
periodic  blocking  of  the  tube.  The  r-f  voltage,  rectified  by  the  grid,  produces  a  bias  voltage 
across  the  grid  leak  sufficient  to  produce  plate  current  cutoff,  and  oscillations  die  out.  The 
charge  on  the  grid  condenser  leaks  off,  the  tube  again  starts  to  oscillate,  and  the  blocking 
cycle  repeats  itself.  The  frequency  of  blocking  depends  partly  on  the  tube  but  chiefly  on 
the  time  constant  of  the  grid  leak  and  condenser.  The  higher  the  frequency  of  blocking, 
the  greater  the  feedback  required  to  produce  the  effect. 

When  the  frequency  of  blocking  is  increased  to  about  the  limit  of  audibility,  by  decreas- 
ing the  values  of  the  grid  leak  and  condenser,  the  circuit  becomes  an  extremely  sensitive 
receiver.  It  is  even  more  sensitive  than  the  regenerative  receiver,  and  much  less  critical 
to  adjust.  The  disadvantage  is  extremely  broad  tuning. 

The  sensitivity  to  weak  signals  is  due  to  the  fact  that  an  oscillator  cannot  start  oscillat- 
ing in  the  absence  of  an  impulse  to  start  it.  Weak  impulses  in  the  form  of  noise  are  always 
present.  It  is  only  necessary  for  the  signal  to  exceed  the  random  noise  in  order  to  control 
the  oscillation.  Since  the  peak  amplitude  of  each  block  of  oscillation  is  very  closely  the 
same  regardless  of  signal  amplitude,  it  is  difficult  to  see  how  audio-frequency  signals  are 
produced.  Probably  the  effect  of  the  signal  is  to  increase  the  frequency  of  blocking  by 


7-118 


VACUUM-TUBE  CIRCUIT  ELEMENTS 


Regeneration 


causing  oscillation  to  start  sooner  each  blocking  cycle.  This  decreases  the  plate  current, 
since  the  tube  is  cut  off  a  greater  percentage  of  the  time.^  Thus  an  amplitude-modulated 
signal  produces  audio-frequency  currents,  of  the  modulation  frequency. 

Superregeneration  Employing  One  Tube  Oscillating  at  Two  Frequencies.     Improved 
results  over  the  above  can  be  obtained  with  the  circuit  of  Fig.  4.    The  tube  of  this  circuit 

oscillates    continuously    at    the 
-  Choke  quench  frequency  of  about  15  kc, 

drawing  grid  current  at  one  point 
in  each  cycle.  When  not  drawing 
grid  current  the  tube  and  circuit 
are  in  a  suitable  condition  to  os- 
cillate at  the  frequency  of  recep- 
tion. However,  when  only  weak 
pulses  or  signals  are  present  to 
trigger  oscillation,  the  voltage 
does  not  have  time  to  build  up 
to  full  amplitude  before  it  is 
quenched  by  grid  current.  Under 
these  conditions  the  amplitude  of 
the  r-f  voltage  on  the  grid,  when 
grid  current  starts,  is  propor- 
tional to  the  actuating 
amplitude. 


Fro.  4.    One-tube  Superregenerative  Receiver 


This  circuit  has  a  better  signal-noise  ratio  than  the  blocking  type,  but  it  is  almost 
equally  broad  in  tuning.  The  chief  application  for  these  circuits  is  for  reception  at  very 
high  frequencies.  For  this  use  the  broadness  of  tuning  is  frequently  an  advantage,  help- 
ing to  find  and  hold  the  signal. 

THE  TUNED  RADIO  -FREQUENCY  RECEIVER.  A  tuned  r-f  receiver  consists  of 
several  stages  of  tuned  r-f  amplification  followed  by  a  detector  and  audio  amplifier. 

Regeneration  in  Multistage  Amplifiers.  In  a  multistage  tuned  r-f  amplifier,  if  capaci- 
tance exists  between  control  grid  and  plate,  regeneration  will  result.  In  fact,  coupling  of 
any  sort  between  any  two  stages  of  an  amplifier  will  result  in  regeneration,  or  oscillation, 
depending  on  the  degree  of  coupling. 

Resistance  Stabilization.  Since  regeneration  is  equivalent  to  adding  negative  resist- 
ance, its  effects  may  be  largely  counterbalanced,  at  a  given  frequency,  by  adding  re- 
sistance to  the  input  circuit.  This  added  resistance  should  not  be  placed  from  grid  to 
filament  if  it  is  desired  to  counteract  regeneration  over  the  whole  tuning  range  with  a 
fixed  value  of  resistance.  The  regeneration  is  much  more  severe  at  the  high-frequency 
end  of  the  tuning  range.  Hence,  it  is  desirable  to  place  the  added  resistance  in  such  a 
position  that  it  will  have  its  greatest  effect  at  high  frequencies.  This  is  accomplished  by 
placing  the  resistance  in  series  with  the  grid. 

^  Tlie  Tuned  R-f  Receiver  with  Resistance  Stabilization.     Figure  5  is  the  schematic 
circuit  diagram  for  a  complete  battery-operated  receiver,  employing  an  untuned  antenna 


Oufput- 
•  B-H33 


Gang  Condensers          Volume  Control       By- Pass  Corvdenser 


FIG.  5.    Tuned  R-f  Receiver,  Resistance  Stabilized 


and  three  r-f  transformers,  with  resistance  stabilization.  Such  a  receiver  would 
e  without  the  presence  of  resistors  n  and  r2.  When  these  resistors  are  given  the 
proper  value  (usually  about  800  ohms)  the  effects  of  regeneration  may  be  approximately 
counterbalanced  over  the  entire  tuning  range.  In  order  to  flatten  the  sensitivity  curve 
the  resistors  are  usually  made  large  enough  to  overcompensate  for  the  regeneration  at 


TYPES  OF  BECEIVERS  7-119 

high  frequencies.    For  this  reason  the  tuning  is  unduly  broad  at  high  frequencies,  in  this 
type  of  receiver. 

The  Neutralized  Receiver.  Another  method  of  eliminating  oscillation  is  by  the  use  of 
neutralization.  (See  p.  7-29.)  Receivers  employing  two  and  three  stages  of  tuned 
amplification  with  capacitance  neutralization  were  quite  popular  before  the  development 
of  the  screen-grid  tube. 

The  Tuned  R-f  Receiver  Employing  Screen-grid  Tubes.  When  screen-grid  tubes  are 
employed  in  a  multistage  tuned  amplifier,  regeneration  of  the  type  discussed  above  is  not 
observed.  The  presence  of  the  screen  grid,  between  control  grid  and  plate,  reduces  the  grid 
plate  capacitance  to  such  a  low  value  that  regeneration  is  appreciable  only  when  the  stage 
gain  is  extremely  high. 

Nevertheless,  it  is  necessary  to  take  many  other  precautions  to  avoid  oscillation  if  the 
overall  gain  is  very  great. 

Other  Sources  of  Regeneration.  Coupling  of  any  sort,  between  any  two  stages  of  the 
receiver,  may  give  rise  to  serious  regeneration.  Coupling  between  adjacent  stages  is  not 
as  serious  as  between  circuits  which  are  one  or  more  stages  removed  from  each  other. 
Capacitative,  or  inductive,  coupling  causes  oscillation  with  equal  facility  because  of  the 
change  in  phase  obtainable  by  tuning  the  intervening  circuits.  Incomplete  shielding  of 
grid  and  plate  leads  is  one  of  the  most  prevalent  sources  of  regeneration. 

When  the  overall  gain  is  high,  the  mutual  inductance  of  the  various  sections  of  the 
gang  tuning  condenser  becomes  troublesome.  Owing  to  the  use  of  a  common  rotor  shaft 
this  coupling  cannot  be  completely  eliminated.  It  can  be  reduced  to  a  satisfactorily  low 
value  by  careful  design.  It  is  general  practice  to  use  several  wiping  contacts  on  the  rotor 
shaft.  The  ground  leads  from  the  tuning  inductances  are  brought  separately  to  different 
terminals  on  the  wiping  contacts,  to  avoid  the  coupling  of  a  common  ground  lead. 

Objectionable  coupling  is  often  caused  by  the  use  of  common  voltage  supplies  for  the 
cathode,  screen,  or  plate  circuits.  This  trouble  can  be  eliminated  by  the  use  of  small 
decoupling  resistors  in  series  with  the  voltage  supply  leads  for  each  tube  and  with  separate 
by-pass  condensers. 

THE  SUPERHETERODYNE  RECEIVER.  The  tuned  r-f  receiver  requires  extreme 
care  to  avoid  oscillation,  because  of  the  high  overall  gain  required  at  radio  frequency. 
To  avoid  this  the  superheterodyne  type  of  receiver,  in  which  unduly  high  gain  is  not 
required  at  any  frequency,  is  used.  Part  of  the  required  amplification  is  obtained  at  the 
radio  frequency  and  part  at  an  intermediate  frequency.  This  makes  stabilization  rela- 
tively easy. 

The  essential  idea  of  the  superheterodyne  receiver  is  to  amplify  all  signals  at  the  same 
fixed  frequency.  The  essential  component  parts  are  the  preselector,  the  frequency  con- 
verter, the  intermediate-frequency  amplifier,  the  audio  amplifier,  and  the  loud  speaker. 
A  typical  receiver  of  this  type  is  shown  in  Fig.  6. 

The  Preselector.  The  preselector  consists  of  an  antenna  input  circuit  with,  or  without, 
one  or  more  tuned  r-f  amplifier  stages. 

The  operating  characteristics  of  the  preselector  are  identical  with  those  of  corresponding 
units  in  a  tuned  r-f  receiver. 

The  preselector  assists  in  producing  discrimination  against  signals  on  adjacent  fre- 
quencies, but  the  intermediate-frequency  amplifier  is  so  much  more  effective  for  this 
purpose  that  the  use  of  the  preselector  is  not  warranted  for  this  alone.  Its  essential  func- 
tion is  the  elimination  of  undesired  responses  at  frequencies  widely  different  from  that  of 
resonance. 

The  Frequency  Converter.  Frequency  conversion  is  obtained  by  the  use  of  the  first 
detector  and  oscillator.  In  most  receivers,  the  oscillator  operates  at  a  frequency  higher 
than  the  signal  frequency.  The  difference  in  frequency  is  the  intermediate  frequency. 
Voltage  from  the  oscillator  and  from  the  signal  is  fed  to  the  first  detector,  and  the  output 
is  amplified  in  the  i-f  amplifier.  Previous  to  1933,  a  majority  of  receivers  employed  sep- 
arate tubes  for  the  oscillator  and  first  detector,  usually  using  a  circuit  in  which  the  oscilla- 
tor voltage  was  fed  to  the  first  detector  cathode.  A  majority  of  modern  receivers  employ 
a  single  tube  in  which  the  two  functions  are  combined.  An  example  of  this  type  of  tube  is 
the  2A7,  used  in  the  circuit  of  Fig.  6. 

Combined  First  Detector  and  Oscillator.  In  the  2A7  tube  the  first  two  grids,  adjacent 
to  the  cathode,  comprise  the  oscillator  elements.  The  voltage  fluctuations  of  the  first  grid 
control  the  electron  stream,  not  only  to  the  oscillator  plate  (called  second  grid  for  con- 
venience although  it  includes  only  two  vertical  rods) ,  but  also  that  to  the  remainder  of  the 
elements.  The  current  arriving  at  the  output  plate  of  the  tube  consists  of  pulses,  at  the 
frequency  of  the  oscillator.  The  amplitude  of  these  pulses  may  be  varied  by  the  control 
grid.  The  remaining  grid,  or  screen,  acts  as  a  shield  between  the  oscillator  elements  and 
the  control  grid  and  output  plate.  It  also  screens  the  control  grid  from  the  output  plate. 


7-120 


VACUUM-TUBE   CIRCUIT  ELEMENTS 


TYPES  OP  RECBIVEES 


7-121 


The  component  at  the  difference  between  oscillator  and  signal  frequency  corresponds 
to  the  intermediate  frequency;  it  is  selected  and  amplified  by  the  i-f  amplifier. 

Tracking.  One  of  the  most  important  problems  of  superheterodyne  design  is  tuning 
the  oscillator  and  the  preselectors  with  a  gang  tuning  condenser.  The  problem  is  to  main- 
tain the  oscillator  at  a  uniformly  higher  frequency  than  the  preselector,  as  the  preselector 
is  tuned  over  the  frequency  band.  The  frequency  difference  must  remain  equal  to  the 
intermediate  frequency.  Since  the  oscillator  is  operated  at  a  higher  frequency  than  the 
preselector,  it  has  a  tendency  to  change  frequency  too  rapidly. 

One  method  of  correcting  this  is  to  specially  shape  the  rotor  plate  of  the  tuning  con- 
denser which  is  used  in  the  oscillator  section  of  the  gang  tuning  condenser.  Another 
method  is  by  means  of  fixed  condensers  in  series  and  shunt  with  the  oscillator  tuning 
condenser.  These  reduce  the  rate  of  change  of  frequency.  When  the  two  auxiliary  con- 
densers and  the  oscillator  inductance  have  the  proper  value,  the  oscillator  frequency 
deviates  only  slightly  from  that  desired,  as  shown  in  Fig.  7. 


w 


<Sr 
_$_ 


V 


14.00 


1800 


l&OO 


\ 


\ 


\ 


1100          1000  900  800 

ri^yca.esjier  Second 
FIG.  7.     Oscillator  Tracking  under  Ideal  Conditions 


700 


600  550 


Figure  8  is  a  curve  which  is  useful  in  deterrnining  the  proper  values  for  the  oscillator 
inductance  and  the  capacitances  to  obtain  the  best  tracking  with  the  preselector.  In  the 
curve:  a  is  the  ratio  of  oscillator  tuning  inductance  to  tbe  secondary  inductance  of  the 
r-f  transformer;  Ca  is  the  value  of  the  oscillator  series  condenser;  C/  is  the  difference  be- 
tween the  oscillator  trimmer  capacitance  and  the  r-f  transformer  trimmer  capacitance. 

Values  of  Cs,  Cf,  and  a  are  plotted  against  intermediate  frequency.  The  lower  abscissa 
scale  is  used,  assuming  that  the  range  to  be  covered  is  550  to  1500  kc.  For  other  ranges  the 
upper  abscissa  scale  should  be  used.  The  curve  is  for  the  condition  of  400  ju/zf  total  circuit 
capacitance  at  the  low-frequency  end  of  the  range.  If  the  maximum  capacitance  is 
changed,  Cs  and  Cf  change  in  the  same  ratio,  while  a  remains  unchanged. 

The  Intermediate-frequency  Amplifier.  The  i-f  amplifier  consists  of  one  or  more  stages 
of  amplification  following  the  first  detector.  The  tuning  is  fixed  at  the  intermediate 
frequency.  Usually  two  coupled  circuits  are  used  in  each  i-f  transformer.  The  frequency 
chosen  usually  lies  between  100  and  500  kc  per  sec.  High  gain  and  good  selectivity  are 
easily  obtained  at  these  frequencies,  particularly  towards  the  lower  of  the  two  values. 

The  Diode  Pentode  Tube.  In  the  circuit  of  Fig.  6,  the  fourth  tube  is  a  combination 
of  a  diode  and  a  pentode  in  a  common  envelope.  The  diode  and  pentode  together  serve  as 


7-122 


VACUUM-TUBE   CIRCUIT  ELEMENTS 


detector  and  audio  amplifier.  This  combination  tube  may  replace  two  separate  tubes  in 
all  circuits  except  those  requiring  different  d-c  cathode  potentials. 

Undesired  Responses.  Although  the  superheterodyne  has  many  advantages,  it  is  sub- 
Sect  to  a  number  of  undesired  responses  and  interfering  beat  notes  which  do  not  occur 
in  a  tuned  r-f  receiver.  However,  careful  design  minimizes  these  difficulties. 

The  Image  Response.  The  most  important  undesired  response  in  a  superheterodyne 
is  known  as  the  "image."  As  explained  above,  the  intermediate  frequency  is  the  difference 
between  the  signal  frequency  and  the  oscillator  frequency.  The  oscillator  is  operated 
above  the  signal  frequency.  However,  i-f  signals  are  produced  equally,  well  by  a  beat 


Intermediate  Frequency 


10.000   1.0  H 


1 


CO  _ 


3.000  0.1     10 


3.0 


100  .10       1 


20         3O  50       70       100  200       30O          500     700     1000  2000      3000 

Lniarmediaie  Frequency  m  Kc- 

PIG.  S.    Proper  Values  of  Oscillator  Inductance  and  Capacitances  for  Best  Tracking  with  the  Preselector 


between  the  oscillator  and  a  signal  which  is  above  the  oscillator  in  frequency.  The  first 
detector  is  equally  responsive  to  signals  at  either  of  these  two  frequencies.  The  only 
means  of  selecting  the  desired  (lower  frequency)  of  these  two  response  points  and  attenu- 
ating the  other  is  by  means  of  the  preselector.  The  higher,  the  frequency  of  the  interme- 
diate amplifier,  the  greater  the  frequency  separation  of  the  desired  signal  and  the  image 
response;  hence,  the  image  response  ratio  is  greater  for  a  high  intermediate  frequency. 
With  an  intermediate  frequency  of  175  kcr  the  ratio  of  the  sensitivity  at  the  desired  response 
to  that  at  the  undesired  response  can  be  made  about  1000  at  the  high-frequency  end  of  the 
broadcast  range  and  about  10,000  at  the  low-frequency  end.  Although  higher  intermediate 
frequencies  give  higher  image  response  ratios,  other  difficulties  may  develop  from  their 
use,  as  described  below. 

Harmonics  of  the  Intermediate  Frequency.  Another  source  of  difficulty  which  may 
be  present  in  the  superheterodyne  is  a  beat  note  which  occurs  when  reception  is  attempted 
afc  a  frecfuency  corresponding  to  a  harmonic  of  the  intermediate  frequency.  The  reason 
to?  tills  beat  note  is  that  the  second  detector  produces  these  i-f  harmonics.  If  a  very  small 
aiaownt  of  coupling  exists  between  the  second  detector  circuit  and  the  antenna,  or  the  r-f 
transforiBex,  the  i-f  harmonic  beats  with  the  incoming  signal  producing  a  disagreeable 
squeal 


TYPES  OF  KECEIVERS 


7-123 


7-124 


VACUUM-TUBE   CIRCUIT  ELEMENTS 


The  higher  the  order  of  the  harmonic  the  less  its  amplitude  in  the  detector  circuit. 
Hence  it  is  easier  to  suppress  the  beat  note  due  to  higher-order  harmonics  than  that  due 
to  the  second  and  third  harmonics.  The  highest  intermediate  frequency  which  can  be 
used  if  the  third  harmonic  is  to  be  kept  outside  of  the  broadcast  frequency  band  is  175 
kc.  This  accounts  for  the  great  popularity  of  this  figure.  When  higher  intermediate 
frequencies  are  used,  such  as  450  kc,  the  severity  of  the  harmonics  in  the  broadcast  band 
is  increased,  but  the  number  of  interference  points  is  reduced  from  five  to  two.  By  very 
careful  shielding,  the  beat  notes  resulting  from  these  harmonics  may  be  almost  completely 
eliminated.  The  greatest  dimculty  is  obtained  with  the  second  harmonic. 

Other  Responses.  The  above  are  the  most  important  of  the  undesired  responses,  but 
many  other  types  occasionally  give  trouble.  For  example,  two  broadcasting  stations  may 
beat  together  to  produce  i-f  signals  independent  of  the  local  oscillator.  Harmonics  of  the 
oscillator  may  beat  with  signals  of  various  frequencies  and  with  their  harmonics,  etc.  A 
good  preselector  is  the  best  insurance  against  all  these. 

ALL-WAVE  RECEIVERS.  The  popularity  of  short  waves  increased  very  rapidly 
during  1933  and  1934.  For  this  reason  most  of  the  commercial  entertainment  receivers 
now  include  provision  for  the  reception  of  frequencies  other  than  the  standard  broadcast 
band.  Many  of  these  receivers  employ  the  name  * 'all-wave,"  but  this  is  a  misnomer  as 
none  of  these  receivers  cover  the  whole  r-f  spectrum.  The  circuit  of  a  typical  receiver  of 
this  kind  is  shown  in  Fig.  9.  The  circuit  is  entirely  conventional,  except  that  the  preselec- 
tor transformers  and  the  oscillator  transformers  may  be  switched  to  any  one  of  the  five 
bands  which  it  covers. 

The  r-f  transformers  for  the  low-frequency  bands  are  purposely  designed  with  restricted 
gain,  so  as  to  maintain  approximately  uniform  sensitivity  on  all  bands.  This  is  necessary, 
because  the  gain  obtainable  in  a  single  stage  is  limited  to  a  low  value  at  high  frequencies. 

RECEPTION  OF  CONTINUOUS  WAVE  CODE  SIGNALS.  In  the  reception  of  un- 
modulated code  signals  it  is  necessary  to  supply  a  local  oscillator  to  produce  an  audible 
beat  note  with  the  incoming  signal.  In  the  regenerative  detector  circuit  of  Fig.  3  it  is  only 
necessary  to  advance  the  tickler  to  a  point  just  beyond  where  oscillation  starts  in  order 
to  receive  this  type  of  signal.  The  regenerative  detector  may  be  preceded  by  one  or  more 
stages  of  tuned  r-f  amplification  to  increase  the  sensitivity. 

The  circuits  of  Figs.  5,  6,  and  9  may  be  used  to  receive  code  signals  by  the  addition  of 
an  external  oscillator.  In  the  tuned  r-f  receiver  the  oscillator  must  be  tuned  to  beat  with 
the  signal  directly.  Hence  it  must  be  tunable  over  the  receiving  frequency  range.  With  ' 
superheterodyne  the  lo<ial  oscillator  may  beat  with  the  signal  at  the  intermediate  frequency. 
Hence  its  tuning  may  be  fixed.  The  oscillator  should  be  coupled  weakly  to  the  detector 
input  circuit. 

TUNING  INDICATORS.  Because  radio  stations  are  not  always  modulating,  it  is 
somewhat  advantageous  to  have  a  visual  indication  of  resonance,  rather  than  to  depend 
on  audio  output  as  a  check  on  the  accuracy  of  timing.  There  are  many  different  methods 
for  accomplishing  this.  One  of  the  simplest  is  to  place  a  milliammeter  in  the  B  supply 
lead  to  one  of  the  r-f  amplifier  tubes  which  is  subjected  to  automatic  volume  control.  The 
stronger  the  signal,  the  higher  the  bias  on  this  tube,  and  the  lower  its  plate  current.  Hence 
the  deflection  of  the  needle  downward  from  its  peak  value  at  no  signal  is  a  good  indication 
of  signal  strength  and  of  the  accuracy  of  tuning. 


1      2         5      10    20       50    100200    500   g      g 

^V.  O         CM 


PIG.  10.     Overload  and  AVC  Curves 


FIDELITY  CHARACTEBISTICS 


7-125 


AUTOMATIC  VOLUME  CONTROL.  The  circuit  connections  for  automatic  volume 
control  are  shown  in  Fig.  6.  The  diode  second  detector  develops  a  d-c  voltage  which  is  so 
connected  as  to  increase  the  bias  on  the  amplifier  tubes  with  increased  signal  strength. 
The  result  of  this  connection  is  that  strong  signals  produce  only  slightly  greater  audio 
response  than  weak  ones.  In  Fig.  10  the  audio  output  of  a  typical  receiver  is  plotted 
against  signal  strength  for  maximum,  and  for  a  reduced  manual  volume  control  setting. 
A  manual  volume  control,  such  as  that  shown  in  the  diode  circuit,  is  necessary  to  adjust 
the  level  of  sound  volume.  After  this  adjustment  is  made  signals  come  in  at  approximately 
the  same  volume.  One  of  the  important  advantages  of  automatic  volume  control  is  a 
reduction  of  the  effect  of  fading  signals. 


27.  FIDELITY  CHARACTERISTICS 

The  circuits  affecting  the  fidelity  characteristics  are:  the  r-f  amplifier,  the  automatic 
volume  control  circuit,  the  audio  amplifier,  the  tone  control,  the  output  transformer,  and 
the  loudspeaker.  The  effect 
of  the  loudspeaker  is  not 
included  in  the  curve  of 
Fig.  11.  Loudspeaker  char- 
acteristics are  discussed  in 
Section  6. 

The  overall  fidelity  curve 
of  Fig.  11  is  the  product  of 
the  fidelity  curves  of  the 
component  parts  of  the  re- 
ceiver. 

EFFECT  OF  R-F  CIR- 
CUITS ON  FIDELITY. 
The  r-f  circuits  affect  the 
fidelity  curve  by  cutting  the 
high-frequency  response. 
The  modulation  on  a  signal 
consists  of  continuous  wave 


M 

O 
0 


a40 

I- 

& 


200    300      500700  1000 
Frequency  In  Cycles  per  Second 

FIG.  11.    Fidelity  Curves  Showing  Effect  of  Tone  Control 


304050    70   100          200   300      5007001000     20003000    4000 

signals  (called  sidebands) 
on  frequencies  adjacent 
to  the  carrier  frequency. 
These  sidebands  differ  in  frequency  from  the  carrier  by  the  value  of  the  modulation  fre- 
quency. The  selectivity  curve  of  the  r-f  and  i-f  amplifier  shows  quite  appreciable  decrease 
at  only  2  or  3  kc  from  resonance.  Hence  the  fidelity  curve  indicates  this  same  decrease  of 
the  high  audio  frequencies. 

THE  EFFECT  OF  THE  AUTOMATIC  VOLUME  CONTROL  ON  THE  FIDELITY. 
The  automatic  volume  control  reduces  the  effects  of  fading,  delivering  to  the  detector  a 
signal  having  only  slight  variations  in  amplitude,  in  spite  of  the  wide  fluctuations  of 
signal  amplitude  on  the  antenna.  In  a  similar  manner  the  amplitude  fluctuations  of  the 
signal,  corresponding  to  low-frequency  modulation,  may  be  almost  completely  wiped  out 
if  the  action  of  the  automatic  volume  control  is  too  fast.  The  action  of  this  circuit  may 
be  slowed  down  by  increasing  the  values  of  the  resistors  and  by-pass  condensers  in  the 
return  leads  of  the  tuning  inductances.  If  the  action  is  too  slow  the  delay  becomes  notice- 
able to  the  ear.  Shocks  of  static  then  blot  out  appreciable  portions  of  the  program,  and  the 
change  in  volume  when  tuning  in  stations  may  be  noticeably  slow.  For  this  reason  the 
time  constant  should  be  low  enough  so  that  there  is  a  small  but  appreciable  effect  on  the 
low-frequency  portion  of  the  fidelity  curve. 

THE  RESISTANCE-COUPLED  AUDIO  AMPLIFIER.  In  Fig.  6  the  diode  section 
of  the  diode-pentode  is  resistance-coupled  to  the  pentode  section.  Also,  the  pentode  is 
resistance-coupled  to  the  output  tube.  Neglecting  the  slight  effect  of  the  grid  leak,  the 
gam  of  a  resistance-coupled  amplifier  is  the  ratio  of  the  voltages  applied  to  the  grids  of  the 
preceding  tube  and  the  following  tube.  It  is  equal  to  E^/JSi  =  j*r/(rp  +  r),  where  p 
is  the  amplification  factor  of  the  tube,  r  is  the  load  resistance,  and  rp  is  the  plate  impedance 
of  the  tube.  Or,  if  the  plate  impedance  is  very  high,  E%/Ei  —  smr,  where  sm  is  the  trans- 
conductance  (or  mutual  conductance)  of  the  tube  at  the  operating  voltages. 

These  formulas  neglect  shunt  capacitance  and  the  coupling  capacitance.  At  high 
frequencies  the  shunt  (plate-filament,  grid-filament,  and  other)  capacitances  affect  the 
r  esult.  The  response  is  attenuated  to  70  per  cent  of  the  mid-range  value,  at  the  frequency 


7-126 


VACUUM-TUBE   CIRCUIT  ELEMENTS 


where  the  shunt  capacitative  reactance  is  equal  to  the  effective  resistance  of  r  and  ry  in 
parallel.  , 

At  low  frequencies  the  coupling  capacitance  reduces  the  gain.  The  response  drops  to 
70  per  cent  at  the  frequency  where  the  reactance  of  the  coupling  capacitor  is  equal  to  the 
resistance  of  the  grid  leak.  The  effect  of  the  diode  resistance-coupling  circuit  may  be 
calculated  in  a  similar  manner. 

TONE  CONTROL.  The  above  paragraphs  neglect  the  effect  on  the  fidelity,  oi  ±C-lb 
and  C-24  (in  Fig.  6),  which  constitute  the  tone  control.  At  the  maximum  setting  of  the 

variable  resistor  the  effect  of  these  two  units 
is  very  slight.  However,  when  the  control 
is  turned  back,  high  frequencies  are  pro- 
gressively attenuated.  A  fidelity  curve  at 
maximum  and  minimum  tone  control  setting 
is  given  in  Fig.  11. 

The  major  use  of  the  tone  control  is  to 
improve  the  apparent  signal  to  noise  ratio. 
Noise  is  usually  uniformly  distributed  over 
the  audio-frequency  spectrum,  while  the 
signal  energy  is  chiefly  contained  in  its 
lower  frequency  components.  Hence  when 
the  tone  control  is  turned  back  the  signal  is 
a  smaller  percentage  than  the 


input 


Output 


FIG.  12.    Compensated  Volume  Control  Circuit 


reduced  by 
noise. 

The  tone  control  may  also  sometimes  be  used  to  improve  faulty  fidelity  in  the  trans- 
mitted signal. 

THE  EFFECT  OF  AUDIO  TRANSFORMERS  ON  THE  FIDELITY.  Audio  trans- 
formers of  either  the  interstage  or  output  type  affect  the  fidelity  by  cutting  both  the 
Mgh-  and  low-frequency  response.  However,  the  degree  may  be  largely  controlled  by 
design.  (See  pp.  6-13  to  6-25.) 

COMPENSATED  VOLUME  CONTROL.  To  the  human  ear  the  apparent  loudness 
of  sound  at  various  frequencies  changes  at  a  different  rate  as  the  amplitude  of  the  sound 
waves  is  changed.  This  makes  it  desirable  to  accentuate  low  frequencies  and  high  fre- 
quencies when  the  volume  control  is  turned  down.  A  circuit  employed  to  accomplish 


orvn 

Percent  of  Response  at,400*v 

5  »  «  ft  g  §  g  } 

fi,O  o  o  <5  ooooc 

1. 
2. 

0. 
0. 
10. 

32 

50 
30 

W. 

w. 

W. 

Pow« 

r  Output  a 

1400 

*v 

3. 

1 

A'~ 

\ 
\ 

> 

/ 

f 

I 

IV 

s 

/ 
/ 

s"~" 

&*5 

^ 

>ft 

*f* 

^ 

*  *£," 

Volurne 

^s 

^< 

^ 

(0       50    70 

100          200              500 

1000       2OOO           5000      10^000 

Frequency  In  Cycles  per  Second 
Fio.  13.     Variation  of  Frequency  Response  with  Setting  of  Compensated  Volume  Control 

this  is  given  in  Fig.  12.  Figure  13  gives  the  fidelity  curve  of  a  receiver  incorporating  this 
artfait.  A  receiver  employing  compensated  volume  control  has  a  more  natural  sound  at 
all  -woiume  levels. 

HOISE  SUPPRESSION.  When  automatic  volume  control  is  employed  with  a  re- 
cersner  of  high  sensitivity,  the  receiver  automatically  goes  to  full  sensitivity  when  tuned 
between  stations.  The  results  are  disagreeably  strong  reproduction  of  the  static  and 
general  isierferenee,  which  is  always  present  in  the  background  at  any  frequency.  In 


RANDOM  NOISE 


7-127 


order  to  make  tuning  more  pleasant,  circuits  of  various  types  have  been  developed  for 
cutting  off  the  audio  amplifier  in  the  absence  of  a  signal  carrier.  One  popular  circuit  for 
this  purpose  is  shown  in  Fig.  14. 

When  no  signal  is  present,  no  current  flows  in  the  diode  circuit.  Hence,  no  bias  voltage 
is  applied  to  the  grid  of  T2  and  maximum  plate  current  flows  through  r2.  The  voltage 
developed  across  r%  biases  V$  to  cutoff  so  that  it  cannot  amplify.  The  audio  signals  from 
interfering  noises  cannot  pass  this  point.  When  an  r-f  signal  is  present,  current  flows  in 
the  diode  circuit  producing  d-c  and  a-c  voltages  across  the  diode  circuit  resistor.  The 


FIG.  14.     Noise  Suppressor  Circuit 

d-c  voltage  biases  Vz  to  cutoff.  There  is  then  no  voltage  drop  across  r%,  so  Vz  operates 
with  normal  bias.  Vz  then  functions  as  an  amplifier  for  the  audio-frequency  voltage 
applied  to  its  grid  through  £4. 

28.  RANDOM  NOISE 

Random  noise  (sometimes  called  fluctuation  noise  or  Johnson  noise)  is  a  fundamental 
form  of  interference  which  prevents  the  satisfactory  reception  of  signals  below  a  certain 
level. 

Random  noise  comes  from  two  sources,  thermal  agitation  in  circuit  resistances,  and 
shot  effect  in  vacuum  tubes.  Thermal-agitation  noise  comes  from  the  random  motion  of 
electrons  in  a  conductor  due  to  its  temperature.  The  open-circuit  rms  noise  voltage  across 
a  resistor  is 

En  =  VlKTr  A/  (1) 

where  K  =  Boltzmann's  constant  =  1.37  X  10"53  joule  per  degree  Kelvin. 
T  —  absolute  temperature  in  degrees  Kelvin. 
r  =  value  of  the  resistance  in  ohms. 
A/  =  the  effective  noise  bandwidth  of  the  instrument  used  to  measure  the  voltage. 

Since  the  effects  of  shot  noise  are  indistinguishable  from  those  of  thermal  noise,  it  is 
customary  to  measure  the  shot  noise  of  a  vacuum  tube  in  terms  of  the  equivalent  noise 
grid  resistance.  This  is  defined  as  the  value  of  external  grid  resistance  required  to  double 
the  noise  power  output  of  the  tube  over  that  with  the  grid  shorted.  The  value  for  a 
triode  is  approximately, 


gm 

where  gm  is  the  trans  conductance  of  the  tube.    The  value  for  a  pentode  is  approximately 
four  times  as  high.* 

Noise  Factor.  The  "available  power"  of  a  signal  generator  is  the  power  delivered  to 
a  load  resistance  under  the  condition  of  an  impedance  match.  Thus  the  available  noise 
power  of  a  resistance  considered  as  a  noise  generator  is: 

-^-  «  KT  A/  (3) 

K  the  resistance  r  is  the  impedance  of  a  receiving  antenna,  and  if  no  other  noise  sources 
existed,  then  a  signal  noise  ratio  of  unity  would  be  obtained  when  the  signal  power  avail- 
able from  the  antenna  was  KT  A/.  This  is  the  (unattainable)  ideal  which  can  never  be 

*  B.  J.  Thompson,  D.  O.  North,  and  W.  A.  Harris,  Fluctuations  in  Space-charge-limited  Currents 
at  Moderately  High  Frequencies,  RCA  Rev.,  VoL  IV,  No.  3  (January  1940). 


7-128  VACUUM-TUBE  CIECUIT  ELEMENTS 

improved  upon.  Actually  other  noise  sources  always  exist,  so  that  the  signal  required 
for  unity  signal  noise  ratio  is  always  greater  than  KT  A/.  The  ratio  by  which  it  is  greater 
is  called  the  noise  factor.  This  ratio  is  usually  expressed  in  decibels.  Below  100  Me  re- 
ceivers can  be  built  that  have  a  noise  factor  only  a  few  decibels  above  thermal.  Above 
600  Me,  10  db  above  thermal  is  considered  good. 

The  test  for  "noise  factor"  is  not  yet  accepted  by  the  Institute  of  Radio  Engineers  as 
a  standard  test,  but  its  use  by  the  armed  services  during  World  War  II  became  so  wide- 
spread that  its  adoption  as  a  standard  seems  inevitable. 

The  Nature  of  Random  Koise.  Random  noise  may  be  considered  to  be  made  up  of 
an  infinite  number  of  sinusoidal  components  of  different  frequency.  The  amplitude  of 
any  single  frequency  component  is  infinitesimal,  but  in  any  finite  bandwidth  the  rms 
voltage  is  proportional  to  the  square  root  of  the  bandwidth  though  independent  of  the 
mean  frequency. 

Distribution  of  Amplitude.  The  actual  voltage  at  any  instant  cannot  be  predicted, 
but  an  accurate  statistical  prediction  can  be  made  of  the  fraction  of  the  time,  taken  over 
a  long  period,  that  the  voltage  will  exceed  any  given  value.  This  fraction  of  the  time  is 
identical  with  the  probability  that  the  given  voltage  V  will  be  exceeded  at  a  given  instant 
and  is  equal  to 


where  E  is  the  rms  voltage  of  the  noise. 

Distribution  of  Envelope  Amplitude  vs.  Time,  If  the  bandwidth  of  the  circuit  pass- 
ing the  noise  is  small  compared  to  the  mean  frequency,  then  the  amplitude  never  changes 
abruptly  from  on©  cycle  to  the  next.  Thus,  in  a  graph  of  the  wave,  if  the  peaks  of  adjacent 
cycles  are  connected  with  a  smooth  line  the  resulting  line  is  the  envelope  of  the  wave  and 
is  a  function  varying  much  more  slowly  than  the  wave  itself.  The  probability  that  the 
envelope  will  exceed  a  certain  value  A  at  any  instant  is 


The  Values  of  Variotis  Averages.f    The  average  absolute  value  of  the  voltage  is: 
F  = 


The  mean  value  of  the  envelope  is:  A  —  1.252.S. 

The  root  mean  square  deviation  of  the  envelope  from  its  mean  value  is: 


Ar  »  <X655#  (6) 

The  root  mean  square  value  of  the  envelope  is:  .Anns 
The  most  probable  value  of  the  envelope  is:  AP  =  ±E. 

BIBLIOGRAPHY 

Bxperimeiatal  Wirdes*  and  Wireless  Engineering, 

Proc.  LR-E. 

Sfcarky,  K.  R^  Radio  Receiver  Design,-    John  Wiley  (1943). 

Terma®,  F.  ^.Radio  Engineering.     McGraw-Hill  (1937). 

Zepler,  E.  E,,  The  Technique  of  Radio  Design.     John  Wiley  (1943). 


RADIO  TRANSMITTERS 

By  J.  E.  Y<mng 

A  radio  transmitter  is  defined  as  a  device  for  producing  r-f  power  for  purposes  of  radio 
1a*ansmission.  It  also  contains  means  of  modulating  or  varying  that  r-f  power,  designated 
as  the  carrier  wave,  in  correspondence  to  the  intelligence  it  is  desired  to  transmit.  Tele- 
vision transmitters  or  others  using  the  pulse  technique,  as  well  as  those  employing  fre- 
$oetxry  modulation,  are  discussed  in  other  sections.  See  Sections  8,  9,  and  20.  This  sec- 
tkm  will  be  concerned  with  broadcast  and  communications  transmitters  employing  ampli- 
t**de  modulation. 

,  "^be  Distribution  of  Amplitude  with  Time  in  Fluctuation  Noise,  Proc.  I.R.E., 
.  oCr-oo  (February  1941). 


INTERMEDIATE-RADIO-FKEQTJENCY  AMPLIFIERS      7-129 

NATIONAL  AND  INTERNATIONAL  REGULATIONS.  Since  radio  communication, 
broadcast  or  point-to-point,  involves  transmission  through  a  common  medium,  it  has 
been  necessary  to  set  up  national  and  international  rules  defining  the  frequencies,  or  chan- 
nels, frequency  tolerance,  and  type  of  emission  for  all  radio  stations.  In  addition,  the 
Federal  Communications  Commission  has  set  up  national  rules  and  standards  governing 
frequency  assignments,  transmitter  power,  time  when  testing  is  permitted,  specifications 
of  performance  regarding  distortion,  noise,  and  fidelity,  grades  of  operators  to  be  em- 
ployed, etc.,  for  all  types  of  radio  transmission.  Other  agencies,  particularly  the  Radio 
Manufacturers  Association,  have  been  active  in  setting  up  recommended  standards  of 
performance  and  uniform  methods  of  writing  specifications  and  testing  for  various  classes 
of  transmitters. 

RADIO  TRANSMITTER— SCOPE.  The  transmitter  is  usually  considered  to  consist 
of  all  audio  equipment  operating  above  standard  telephone-practice  levels,  and  all  r-f 
equipment  from  the  source  of  the  r-f  oscillations  to  the  transmission  line  connecting  the 
transmitter  to  the  antenna.  Within  the  equipment  defined  by  these  limits  is  contained 
sufficient  audio-frequency  amplification  to  raise  the  input  signal  to  a  level  high  enough  to 
perform  the  function  of  modulation,  r-f  amplification  and  multiplication  to  raise  the 
power  level  and  frequency  of  the  r-f  oscillations  to  the  output  level,  and  the  necessary 
power  supplies  for  these  circuits.  The  function  and  application  of  vacuum  tubes  to  these 
elements  of  the  radio  transmitter  are  discussed  in  other  articles  of  this  section;  only  cer- 
tain general  aspects  of  design  and  performance  will  be  covered  here. 

FREQUENCY  CONTROL.  The  requirements  of  frequency  stability  are  virtually 
always  too  severe  to  permit  modulating  (including  c-w  keying)  an  oscillator  used  as  a 
direct  source  of  antenna  power.  The  usual  practice  is  to  use  a  low-power  oscillator,  sep- 
arated from  the  modulated  amplifier  by  a  sufficient  number  of  stages  to  prevent  interac- 
tion caused  by  changes  in  impedance  of  the  modulated  stage  resulting  from  the  processes 
of  modulation  or  other  variations  in  load. 

For  transmitters  operating  on  fixed  frequencies,  quartz  crystals  are  commonly  used  as 
the  frequency-deterrnining  element.  A  frequency  stability  of  10  parts  per  million  is  quite 
easily  achieved  with  a  quartz-crystal-controlled  oscillator  in  which  the  temperature  of 
the  quartz  plate  is  not  permitted  to  vary  more  than  ±1  deg  cent. 

Some  types  of  transmitters,  particularly  those  used  for  military  applications,  operate 
on  frequencies  which  may  frequently  be  changed.  For  these  applications  quartz  crystals 
are  not  practical  and  a  master  oscillator  in  which  the  tuned  circuit  is  the  frequency-de- 
termining element  is  employed.  Such  oscillators,  having  as  much  as  2  :  1  frequency 
range,  may  be  designed  to  have  a  frequency  stability  better  than  1  part  in  10,000  for 
moderate  variations  of  temperature,  humidity,  and  power  supply  voltages. 

OSCILLATOR  POWER.  It  is  possible  to  design  crystal-controlled  oscillators  which 
will  produce  several  hundred  watts  of  output  power;  however,  such  oscillators  are  difficult 
to  adjust  and  somewhat  less  stable  than  those  of  lower  power  output.  It  is  better  design 
practice  to  use  a  crystal  oscillator  which  has  only  a  few  watts  of  output,  followed  by  high- 
gain  shielded-grid  amplifiers,  to  achieve  maximum  frequency  stability,  and,  in  keyed- 
oscillator  circuits,  freedom  from  chirps  and  frequency  creepage.  This  practice  also  per- 
mits the  use  of  small  crystals  mounted  in  compact  holders,  since  the  crystal  dissipates 
very  little  heat. 

29.  mTEMvIEDIATE-RADIO-FREQT3ENCY  AMPLIFIERS 

The  intermediate-r-f  amplifier  stages  perform  the  triple  function  of  increasing  the  power 
level  of  the  frequency-controlled  oscillator,  multiplying  its  frequency,  if  required,  and 
acting  as  a  buffer  between  the  modulated  amplifier  and  the  oscillator.  In  the  interest  of 
simplicity,  intermediate  amplifier  stages  usually  employ  high-gain,  multigrid  tubes  so  that 
it  is  not  uncommon  to  attain  all  three  of  these  objectives  with  one  or  two  intermediate 
amplifier  stages.  In  high-level  modulated  and  telegraph  transmitters  the  intermediate 
amplifiers  are  operated  class  C.  In  low-level  modulated  transmitters  the  intermediate 
amplifier  stages  following  the  modulated  stage  must  reproduce  the  audio-modulated 
envelope  and  must,  therefore,  be  operated  as  class  B  r-f  amplifiers.  (See  p.  7-22.) 

INTERSTAGE  COUPLING  CIRCUITS.  Amplitude-modulated  transmitters  are  rarely 
used  at  frequencies  above  40  Me.  Up  to  this  frequency  no  special  precautions  are  ordi- 
narily required  in  interstage  circuits.  In  general,  it  is  desirable  to  make  the  circuits  as 
simple  as  possible  to  avoid  dangerous  multiple  resonances  and  parasitics.  Typical  coup- 
ling circuits  for  shielded-grid  tubes  are  shown  in  Fig.  1.  Note  that  the  grid  leak  is  not 
by-passed  and  that  grid  and  plate  feed  circuits  are  dissimilar.  It  frequently  happens 
that  the  input  resistance  of  the  driven  amplifier  is  so  low  that  the  r-f  choke  in  series  with 


7-130 


VACUUM-TUBE  CIRCUIT  ELEMENTS 


the  grid  leak  may  be  eliminated  without  appreciably  affecting  the  driver  power  output. 
This  should  always  be  done  where  calculations  so  indicate,  since  a  possible  parasitic  circuit 
is  thereby  eliminated.  For  the  same  reason  it  is  desirable  to  use  the  voltage  obtained  by 
the  now  of  rectified  grid  current  through  a  resistor  as  grid  bias  rather  than  to  obtain  this 
voltage  from  a  separate  source.  Protection  against  excessive  plate  dissipation,  if  the 
excitation  fails,  may  be  obtained  by  the  use  of  sufficient  cathode  bias. 


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Eia.  1.     Interstage  Coupling  Circuits 

CIRCUIT  0,  or  KVA/KW  RATIO.  Unmodulated  intermediate  amplifier  stages  are 
not  critical  with  respect  to  interstage  coupling  circuit  Q,  although  it  is  desirable  that  the 
circuit,  looking  from  the  grid  of  the  driven  tube,  be  of  high  enough  Q  to  insure  sinusoidal 
wave  shape  in  spite  of  the  variable-impedance  characteristic  of  the  grid,  as  it  is  driven 
positive.  Unfortunately,  in  the  grounded  filament  circuits,  the  voltage  relations  are  such 
that  the  driving  tube  puts  energy  into  the  coupling  circuit  almost  180°  away  from  the 
time  when  it  is  absorbed  by  the  grid  of  the  driven  amplifier.  The  energy  storage  must  be 
sufficient  to  take  care  of  this  condition.  The  wave  shape  will  be  adequate  if  the  circuit 
Q  is  greater  than  10,  unless  the  driven  tube  grid  impedance  at  the  peak  of  the  positive 
grid  swing  is  extraordinarily  low.  Unless  the  driving  stage  is  modulated,  even  higher 
values  of  Q  are  useful  and  desirable,  provided  the  concomittant  coupling  circuit  losses  are 
not  thereby  made  excessive, 

HARMONIC  AMPLIFIERS.  Because  of  special  considerations,  such  as  crystal  ac- 
tivity, stability,  etc.,  frequency  multiplication  is  ordinarily  employed  where  the  carrier 
frequency  is  higher  than  10  Me,  The  necessary  multiplication  is  accomplished  in  the  inter- 
mediate amplifier  stages.  The  multiplication  through  any  one  amplifier  stage  may  be 
from  twice  to  as  high  as  five  times.  The  plate  efficiency  of  the  amplifier  tube  is  very 
neiyriy  inversely  proportional  to  the  order  of  multiplication,  if  the  most  favorable  condi- 
tions of  loading  and  open  angle  of  plate  current  are  chosen  for  each  multiplication.  Be- 
cause of  the  relatively  low  efficiency  obtained  when  multiplying  five  times,  this  amount  of 
multiplication  is  rarely  used  in  any  one  stage.  Because  of  the  generally  poorer  plate-circuit 
©ffiekaaey,  multiplication  is  usually  accomplished  at  low  power  level,  followed  by  amplifiers 
tuafed  to  the  output  frequency,  to  boost  the  power  level  as  required. 


POWER  AMPLIFIERS 


7-131 


Optimum  Angle  of  Current  Flow.  The  term  "angle  of  current  flow"  represents  the 
portion  of  the  grid  voltage  cycle  during  which  plate  current  flows,  expressed  in  degrees. 
The  optimum  angle  of  current  flow  depends  on  how  nearly  the  plate-current  plate-voltage 
characteristic  conforms  to  the  3/2  relation,  and  for  a  3/2  characteristic  may  be  shown  to  be 
approximately  130°  to  develop  maximum  second-harmonic  voltage  in  the  plate  circuit, 
85°  for  the  third,  65°  for  the  fourth,  and  50°  for  the  fifth.  The  angle  of  current  flow  is 

given  by  the  relation  cos  —  =  ~ ,  where  6  is  the  angle  of  current  flow  in  degrees 

of  the  excitation  frequency,  Ec  is  the  grid  bias,  E0  the  projected  cutoff  bias,  and  Eg  the 
grid  excitation  voltage.  Negative  bias  voltages  should  be  written  in  as  negative  numbers. 
Calculation  will  show  that  an  open  angle  of  60°  may  be  obtained  with  a  bias  voltage  of 
not  less  than  ten  tunes  cutoff  bias.  As  the  open  angle  is  further  restricted,  the  required 
bias  voltage  increases  rapidly.  It  is  not  usually  economical  to  use  an  open  angle  of  plate 
current  flow  of  less  than  60°  or  to  attempt  to  raise  the  frequency  more  than  four  times 
through  a  single  amplifier  stage.  This  does  not  hold  true  where  the  output  power  required 
is  small,  as  in  frequency-measurement  work,  where  only  milliwatts  of  power  are  usually 
required,  and  where  the  frequency  may  be  multiplied  many  times, 

30.  POWER  AMPLIFIERS 

Power  amplifiers  may  be  either  grid-  or  plate-circuit  modulated.  (See  Modulators, 
pp.  7-73  and  7-74.)  If  the  former,  the  power  amplifier  must  be  so  operated  as  to  reproduce 
accurately  in  its  plate  circuit  an  r-f  envelope  having  the  same  wave  shape  as  that  of  the 
excitation  voltage.  This  may  be  accomplished  by  using  either  a  linear  amplifier  or  a 

Shunt  neutralizing 


HI— 


Plate 
tuning 


Loading 


+dc 
FIG.  2. 


Typical  R-f  Amplifier 


high-efficiency  Doherty  amplifier.  The  latter  circuit  utilizes  linear  amplifiers  in  pairs 
and  provides  for  dynamic  changes  in  drive  and  loading  so  that  the  average  plate-circuit 
efficiency  is  several  times  that  which  may  be  obtained  with  a  conventional  amplifier. 
Two  circuit  paths  are  provided;  in  the  first,  the  amplifier  tube  is  adjusted  to  operate  at 
the  upper  knee  of  its  dynamic  characteristic.  Between  its  plate  circuit  and  the  load  is 
connected  a  90°  phase-shifting  network.  The  second  tube  is  biased  so  that  its  plate  cur- 
rent is  almost  zero  under  carrier  conditions.  As  the  drive  increases  during  the  positive 
excursions  of  the  modulation  cycle,  the  plate  current  of  this  tube,  and  its  power  output, 
increase  in  the  same  manner  as  an  overbiased  linear  amplifier.  The  effective  load  resist- 
ance seen  at  the  output  of  the  90°  network  connected  in  the  plate  circuit  of  the  first  tube 
increases  as  the  second  tube  supplies  power  to  the  load.  Because  of  the  familiar  impedance 
characteristic  of  the  90°  network,  this  results  in  a  reduction  in  resistance  at  the  input 
terminals  of  this  network.  Therefore,  the  effective  load  resistance  of  the  first  amplifier  is 
reduced  and  its  power  output  is  proportionately  increased.  Thus,  at  the  peak  of  the 
positive  excursion  of  the  driving  voltage,  the  power  output  of  the  first  tube  is  doubled, 
because  of  its  increased  loading,  and  the  output  of  the  second  tube  is  increased  in  the 
familiar  fashion  of  a  linear  amplifier,  so  that  it  is  also  twice  the  carrier  output.  The  total 
power  output  of  the  amplifier  is  thus  four  times  the  carrier  level,  fulfilling  the  condition 
for  100  per  cent  upward  modulation.  On  the  downward  excursion  of  the  modulation 
cycle  the  second  amplifier  is  biased  off  completely,  and  the  first  amplifier  functions  as  a 
conventional  linear  amplifier. 


7-132 


VACUUM-TUBE    CIRCUIT  ELEMENTS 


The  envelope  wave-form  distortion  resulting  from  the  use  of  the  Doherty  amplifier  is 
usually  too  great  to  be  tolerable  in  high-fidelity  systems.  This  defect  may  be  remedied 
by  applying  overall  feedback  from  the  output  terminals  of  the  amplifier,  through  an  r-f 
rectifier  back  into  the  audio-frequency  amplifier  circuits  at  some  convenient  point. 
>  Amplifiers  employing  plate-circuit  modulation  are  termed  "high  level."  In  practice 
the  amplifier  is  operated  class  C,  and  the  energy  to  perform  the  modulation  function  is 
provided  by  a  class  B  modulator.  (See  Modulators,  p.  7-74.)  _  High-level  amplifier 
output  circuits  must  provide  the  necessary  impedance  transformation  between  the  plate 
circuit  of  the  tubes  and  the  load  and  in  addition  must  provide  sufficient  r-f  harmonic  at- 
tenuation to  prevent  excessive  harmonic  radiation.  In  practice,  the  Q  of  the  output  tank 
circuit  is  generally  made  quite  low,  from  4  to  8  for  single-ended  amplifiers,  and  additional 
harmonic  attenuation  is  obtained,  if  necessary,  by  a  low-pass  filter  inserted  between  the 
terminals  of  the  amplifier  and  its  load.  A  typical  amplifier  circuit  employing  shunt  neu- 
tralization is  shown  in  Fig.  2. 

The  high-level,  modulated  amplifier  has  the  advantage  that  it  is  simple  to  adjust  and 
uncritical  in  its  operation.  In  addition,  its  quiescent  (carrier  level)  efficiency  is  very  high. 
Since  the  average  percentage  of  modulation  rarely  exceeds  15  per  cent  in  broadcast 
operation,  the  quiescent  efficiency  is  a  most  important  consideration  in  the  economics  of 
transmitter  operation. 

R-F  HARMONIC  RADIATION".  Under  the  present  conditions  of  crowded  frequency 
assignments  throughout  the  r-f  spectrum,  the  first  few  multiples  of  the  transmitter  fre- 
quency— the  frequencies  that  contain  most  of  the  harmonic  energy — will  often  interfere 


»  furvdemental  frequency 

harmonic  frequency 
FIG.  3.    Harmonic  Attenuator 

with  some  other  radio  service.  Thus,  because  of  a  particular  interference  problem,  one 
harmonic  often  must  be  reduced  far  more  than  the  rest.  The  problem  should  first  be 
analyzed,  by  making  measurements  to  ascertain  definitely  that  the  interfering  signal  re- 
sults from  direct  radiation  from  the  station  or  its  antenna.  It  frequently  happens  that  a 
wire  or  metal  surface  located  near  the  point  of  interference,  or  even  the  receiving  antenna 
itself,  is  picking  up  energy  from  the  transmitter  and  through  the  agency  of  an  oxide- 
covered  joint  is  itself  producing  the  harmonic  signal.  This  may  be  checked  by  field-in- 
tensity measurements  made  at  a  series  of  points  on  a  line  from  the  transmitter  to  the 
point  of  interference.  Peaks  in  the  harmonic  signal  intensity  will  be  observed  in  the  vicinity 
of  such  harmonic  generators.  The  remedy  is  obvious  but  not  always  simple.  Very  often, 
however,  a  single  guywire  or  other  metal  object  having  dimensions  comparable  to  the 
wavelength  in  question  will  be  found  to  be  the  source  of  the  interfering  signal. 

Tbe  field-intensity  meter  may  also  be  used  to  determine  whether  the  interfering  har- 
monic is  radiated  from  the  transmitter  directly  or  from  the  transmitting  antenna  if  the 
two  are  sufficiently  separated  so  that  a  directional  fix  may  be  obtained.  If  the  radiation 
ajppears  to  be  from  ihe  antenna,  the  signal  energy  may  be  carried  by  the  transmission 
tee  from  the  transmitter  to  the  antenna,  or  it  may  again  result  from  rectification  either 
«fc  m  oxidized  metal  joint  or  at  a  tube  rectifier,  such  as  is  used  for  remote  antenna  cur- 
raisfc  leading  or  feedback.  If  the  antenna  rectifier  is  the  offender  the  trouble  may  be  cured 
gr  eonaeetog  a  Jew-pass  filter,  designed  to  cut  off  just  above  the  transmitter  frequency, 
^reea  t&e  rectifier  and  the  point  where  it  picks  up  its  energy  from  the  transmitter 
output. 


POWER  AMPLIFIERS  7-133 

If  the  interfering  signal  emanates  from  the  transmitter  and  is  carried  to  the  antenna  by 
the  transmission  line,  a  trap,  consisting  of  a  circuit  parallel-resonant  to  the  harmonic,  in 
series  with  the  transmission  line,  and  a  circuit  series-resonant  to  the  harmonic  in  shunt 
with  the  line,  will  usually  result  in  sufficient  attenuation  to  eliminate  the  interference. 
Such  a  circuit  may  be  designed  so  that  its  insertion  will  not  affect  the  transmitter  per- 
formance at  its  fundamental  frequency,  and  it  is  preferably  constructed  so  that  the  tuning 
of  at  least  the  shunt  arm  is  variable,  to  permit  exact  adjustment  to  the  harmonic  fre- 
quency. Figure  3  shows  one  typical  circuit. 

If  a  balanced  transmission  line  is  used,  the  trap  circuits  should  be  symmetrical,  with 
their  center  taps  grounded  to  prevent  in-phase  transmission  of  the  harmonic  along  the 
transmission  line  wires. 

If  the  directional  fix  indicates  that  the  transmitter  itself  is  radiating  the  interfering 
signal,  it  may  be  necessary  to  resort  to  complete  shielding  to  eliminate  the  trouble.  More 
frequently,  sufficient  improvement  will  be  obtained  by  other  minor  design  changes,  such 
as  better  grounding,  more  direct  return  of  plate  tank  lead  to  filament  of  power  amplifier 
or  elimination  of  stray  capacitance  coupling  between  plate  and  output  connection. 

NEGATIVE  FEEDBACK.  The  distortion,  noise,  and  frequency  characteristics  of  a 
telephone  transmitter  may  be  considerably  improved  by  the  application  of  negative  feed- 
back over  all  or  part  of  the  system.  Two  general  methods  of  use  of  negative  feedback 
are  common.  In  transmitters  using  low-level  modulation,  followed  by  linear  amplifiers, 
feedback  voltage  is  derived  from  an  r-f  rectifier  coupled  to  the  output  of  the  power  am- 
plifier. This  signal,  of  the  proper  amplitude  to  secure  the  desired  feedback,  is  introduced 
into  one  of  the  audio-frequency  amplifier  stages.  Since  the  r-f  rectifier  is  in  the  0  loop  of 
the  feedback  circuit,  noise  or  distortion  generated  hi  it  will  appear  in  the  ouput  of  the 
transmitter.  It  must,  therefore,  be  carefully  designed  to  reproduce  accurately  the  audio 
envelope  of  the  output  r-f  signal. 

One  difficulty  frequently  encountered  in  the  application  of  this  type  of  feedback  to 
broadcast  transmitters  is  that  the  transmitter  antenna  also  functions  as  a  receiving  an- 
tenna for  signals  from  other  broadcast  stations.  Sufficient  voltage  may  be  developed 
in  the  circuit  to  which  the  feedback  rectifier  is  connected  by  another  broadcasting  station, 
located  in  the  immediate  vicinity,  to  generate  a  series  of  cross-modulation  products  in 
the  rectifier.  Any  of  these  new  frequencies  lying  within  the  pass  band  of  the  a  and  |8 
loops  of  the  audio  system  will  produce  sidebands  which  will  be  radiated  by  the  antenna 
and  may  cause  serious  interference.  Such  effects  may  be  greatly  reduced  by  installing 
a  trap  tuned  to  the  frequency  of  the  other  station  in  the  antenna  system  beyond  the  pick- 
up point  for  the  feedback  rectifier.  Care  must  be  used  in  designing  such  traps  that  they 
do  not  cause  sufficient  phase  shift  to  alter  the  phase-shift  attenuation  characteristic  of  the 
transmitter  seriously  or  instability  may  result. 

The  negative  feedback  will  correct  for  distortion  and  noise,  but  only  so  long  as  there  is 
reserve  capability  in  the  system  to  effect  the  correction.  For  example,  it  cannot  correct 
for  the  distortion  arising  from  overmodulation  and,  in  fact,  by  introducing  high-order 
harmonics  back  into  the  circuit  may  actually  increase  the  distortion  if  the  system  capabil- 
ities are  exceeded. 

Since  it  is  possible  to  keep  the  distortion  of  high-level  modulated  amplifiers  down  to  a 
fraction  of  a  per  cent  by  the  methods  discussed  under  Modulators,  p.  7-74,  negative 
feedback  is  usually  applied  over  the  audio  system,  only,  in  this  type  of  transmitter.  This 
eliminates  the  need  for  the  r-f  rectifier  and  largely  eliminates  the  effect  of  the  output 
amplifier  tank  and  load  circuits  on  the  performance  of  the  feedback  loop.  The  feedback 
voltage  is  usually  obtained  from  a  voltage  divider,  connected  between  the  plate  of  each 
modulator  tube  and  ground.  The  divider  consists  of  capacitors,  shunted  by  resistors. 
These  elements  are  proportioned  so  that  their  impedances  are  equal  at  a  frequency  of 
approximately  100  cycles  per  second.  The  advantage  of  this  type  of  divider  over  simple 
resistances  is  mainly  that  the  operation  of  the  circuit  is  unaffected  by  the  capacitance  of 
the  leads  connecting  the  network  to  the  low-level  audio-amplifier  stage.  Some  improve- 
ment in  the  phase  characteristic  is  also  secured. 

The  use  of  feedback  permits  high-efficiency  linear  amplifiers  to  be  used  in  services  such 
as  broadcasting,  where  low  distortion  and  low  noise  level  are  essential.  In  high-level 
transmitters,  feedback,  applied  over  the  audio  system,  including  the  modulator,  effects  a 
large  reduction  in  noise  and  distortion  and  reduces  performance  variations  due  to  non- 
uniformity  of  parts  and  tubes.  It  also  improves  the  modulator  efficiency,  permitting  the 
no-signal  plate  current  to  be  reduced  substantially  to  zero. 


7-134  VACUUM-TUBE   CIRCUIT  ELEMENTS 

31.  AUDIO  AMPLIFIERS 

Broadcast  transmitters  are,  by  industry  agreement,  provided  with  enough  audio  gain  to 
produce  100  per  cent  sine-wave  modulation  with  an  audio  input  level  of  +10  db  (1  milli- 
watt reference  level)  at  an  impedance  of  600/150  ohms.  Special-purpose  transmitters 
may  include  more  gain,  although  it  is  usually  advisable  to  limit  the  gain  so  an  input  signal 
of  at  least  -  20  db  is  required.  If  enough  gain  is  to  be  included  in  a  transmitter  to  accept 
a  lower  signal  level,  special  shielding  and  filtering  precautions  are  advisable  to  prevent 
objectionable  feedback. 

Limiting  amplifiers  are  generally  used.  A  limiting  stage  may  be  designed  into  the  audio 
system  of  the  transmitter,  or  an  external  line  amplifier  of  the  limiting-compressing  type 
may  be  used.  Even  in  the  latter  case  it  is  advisable  to  design  one  of  the  transmitter  audio 
stages  so  that  positive  modulation  greater  than  125  per  cent  is  impossible,  regardless  of 
the  input  signal  level.  Owing  to  switching  transients  or  accidents,  input  levels  10  to  20  db 
above  100  per  cent  modulation  are  bound  to  occur,  and,  unless  they  are  prevented  from 
reaching  the  high-power  amplifier  by  an  absolute  safeguard  (such  as  driving  the  plate 
current  of  an  audio  stage  to  cutoff),  serious  damage  may  result. 

Many  special  features  are  incorporated  in  audio  systems  to  meet  the  requirements  of 
particular  services.  Some  of  these  are:  automatic  gain  control,  for  circuits  subject  to 
wide  variations  in  level;  pre-emphasis  of  high  audio  frequencies,  used  on  conjunction  with 
complementary  de-emphasis  in  the  receiver  to  reduce  noise;  and  scramblers  which  invert 
or  otherwise  distort  speech,  making  it  unintelligible  unless  a  receiver  provided  with  a  com- 
plementary restorer  is  used. 

32.  TELEGRAPH  TRANSMITTERS 

For  point-to-point  transmission,  telegraph  code  signals  are  often  employed,  since  it  is 
generally  possible  to  secure  100  per  cent  transmission  with  a  much  weaker  signal  when 
telegraph  is  used  instead  of  voice.  Transmission  may  be  accomplished  either  by  tone 
modulation,  which  directly  replaces  the  voice  transmission,  by  keying  the  carrier  wave 
directly  (usually  by  overbiasing  one  of  the  intermediate  amplifiers),  or  by  shifting  the 
frequency  of  the  carrier.  The  last  method  may  be  accomplished  by  shifting  the  carrier 
a  few  hundred  cycles  back  and  forth  at  an  audio  rate  and  keying  the  audio  signal  or  by 
shifting  the  carrier  between  two  fixed  frequencies,  several  hundred  cycles  apart,  one  cor- 
responding to  the  "mark"  and  the  other  the  * 'space.'*  Both  these  systems  are  akin  to 
frequency  modulation  and  provide  some  advantage  in  noise  suppression  and  diversity 
effect  without  the  complication  of  diversity  receivers. 

33,  INSTALLATION  OF  RADIO  TRANSMITTERS 

TRANSMITTER  TESTING.  A  great  deal  of  test  equipment  is  now  available,  making 
it  possible  to  examine  conveniently  all  phases  of  the  performance  of  a  transmitter.  Tests 
generally  performed  are  as  follows: 

1.  Circuit  check— by  means  of  click  meter,  ohmmeter,  etc.,  to  be  sure  that  wiring  has 
been  done  in  conformance  with  wiring  diagrams. 

2.  Control  circuit  check.    Without  tubes  in  any  circuits,  check  the  functioning  of  line 
switches,  starters,  time  delay  relays,  overload  relays,  interlocks,  and  circuit  breakers. 
No-load  voltages  of  filament  transformers  may  be  checked  at  this  time. 

3.  Transformer  voltages.     With  filament  circuits  energized  and  tubes  in  place,  check 
filament  voltages,  transformer  tap  settings,  range  of  primary  voltage  control,  functioning 
of  air  blower  and/or  water-cooling  system. 

4.  Operational  check.    With  all  circuits  tuned  and  functioning  normally,  check  voltages 
and  currents  on  all  elements  of  all  tubes.    Check  for  instability  and  parasitic  oscillation. 

5.  Modulation  characteristics.    Under  normal  operating  conditions,  check  audio  har- 
monic distortion  at  various  modulation  levels  and  at  modulating  frequencies  throughout 
the  normal  frequency-transmission  range.     The  same  test  equipment  is  required  for 
s*e*K»rement  of  the  frequency  characteristic,  and  residual  noise  level,  so  these  tests  should 
be  performed  at  the  same  time.    Figure  4  shows  the  equipment  set-up  required  for  this 
test. 

&  IWer  output.  This  test  should  be  performed  using  loads  representing  the  high- 
aad  low-knut  load  impedances  which  the  transmitter  may  be  required  to  feed  To  obtain 
»ee«rafce  power  checks,  the  load  resistance  should  be  accurately  measured  and  the  load 


INSTALLATION   OF  RADIO  TRANSMITTERS  7-135 


current  determined  by  means  of  an  accurate  r-f  ammeter;  or  the  calorimetric  method  of 
power  measurement  may  be  used,  in  which  the  increase  in  heat  content  of  water  used  to 
cool  the  dummy  load  is  measured.  The  temperature  rise  of  the  water  in  degrees  centigrade 
multiplied  by  the  number  of  gallons  flowing  per  minute  multiplied  by  the  constant  264 
will  be  equal  to  the  number  of  watts  of  power  dissipated  in  the  load. 

7.  Heat  run.  The  transmitter  should  be  run  under  conditions  of  normal  operation  for 
a  period  long  enough  for  all  components  to  reach  approximately  constant  temperature. 
The  procedure  set  up  by  the  RMA  committee  on  amplitude-modulated  broadcast  trans- 
mitters is  a  good  example  of  how  such  a  test  should  be  run: 

"The  transmitter  shall  be  operated  (at  rated  power  output)  .  . .  long  enough  for  all 
components  to  attain  temperature  stability;  that  is,  until  the  hourly  increment  does  not 
exceed  5  per  cent  of  the  total  change  under  the  following  conditions: . .  .  The  carrier 


Measure  modulation 
capability  and  carrier  shift 


r— 

1  Measure  distortion 
frequency  characteristic 


residual  hum. 
FIG.  4.    Modulation  Characteristics  Measurements 

should  be  continuously  modulated  by  a  1000-cycle  sine  wave.  The  audio  input  level  to 
the  transmitter  shall  be  approximately  10  db  below  that  corresponding  to  100  per  cent 
modulation  for  not  more  than  98  per  cent  of  the  duration  of  the  run  and  not  less  than 
10  db  above  that  corresponding  to  100  per  cent  modulation  for  the  remainder  of  the  time."  * 
Temperature  rise  measurements  of  motor  or  generator  windings,  transformer  windings, 
etc.,  should  be  made  by  the  hot-cold  resistance  method,  in  which  the  resistance  and  tem- 
perature of  the  winding  are  measured  after  the  unit  has  been  out  of  operation  long  enough 
to  reach  a  uniform,  stable  temperature.  The  resistance  at  the  end  of  the  heat  run  is  then 
measured,  and  the  rise  in  temperature  is  computed  by  the  formula: 

_  234fi2 


TI  =  temperature  at  which  cold  resistance  was  measured. 
Tz  —  ambient  temperature  at  end  of  heat  run. 
RI  =  resistance  at  TI. 
JKs  =  resistance  at  end  of  heat  run. 
T  «  rise  of  winding  temperature  above  final  ambient. 

8.  Harmonic  radiation.  If  the  transmitter  is  tested  using  a  dummy  load,  it  is  possible 
to  make  a  rough,  check  on  harmonic  intensity  by  coupling  a  field-intensity  meter  to  the 
load  through  an  attenuator,  shielding  the  field-intensity  meter  so  that  all  its  pick-up  is 
derived  from  the  load.  However,  since  dummy  loads  rarely  ever  even  approximate  the 
impedance  characteristic  of  an  actual  radiating  antenna  system,  such  tests  are  of  little 
quantitative  value,  and  more  satisfactory  results  will  be  secured  if  these  measurements 
can  be  made  on  an  actual  field  installation. 

*  See  Radio  Manufacturers  Association,  Engineering  Department,  Standards  Proposal  172,  Elec- 
trical Performance  Standards  for  Standard  Broadcast  Transmitters. 


7-136  VACUUM-TUBE   CIRCUIT  ELEMENTS 

9.  Incidental  phase  modulation.    Instruments  which  will  measure  the  degree  of  phase 
modulation  at  frequencies  below  40  Me  are  seldom  available.    The  development  of  com- 
mercial equipment  operating  in  the  88-108  Me  band  for  the  purpose  of  measuring  the 
percentage  of  modulation  of  f-m  broadcast  stations,  however,  makes  a  handy  tool  availa- 
ble for  this  purpose.    The  signal  to  be  measured  is  multiplied  up  to  a  frequency  within 
the  range  of  the  meter  by  low-power  harmonic  amplifiers,  and,  since  the  meter  is  cali- 
brated in  terms  of  a  75-kc  swing,  the  actual  swing  may  be  measured,  and^then  converted 
into  degrees  of  phase  modulation,  and  divided  by  the  multiplication  ratio  to  determine 
the  actual  amount  of  phase  modulation  at  the  transmitter  carrier  frequency. 

10.  Telegraph  transmitter  tests.     In  addition  to  the  above  tests,  measurements  are 
made  on  the  character  of  the  keyed  pulses  of  telegraph  transmitters.    The  shape  of  the 
keying  wave,  build-up,  transients,  breaks,  etc.,  may  be  observed  or  photographed  on  the 
face  of  a  cathode-ray  tube  which  has  one  pair  of  plates  connected  to  a  sample  of  the  r-f 
output  of  the  transmitter,  and  the  other  pair  excited  by  a  sweep  voltage  which  is  prefer- 
ably adjusted  to  synchronize  the  picture  of  the  keyed  pulse  on  the  screen.    The  mark-to- 
space  ratio  may  be  calculated  by  measuring  the  d-c  current  output  of  a  rectifier  energized 
from  the  r-f  output  of  the  transmitter.    The  percentage  of  mark-to-carrier  is  given  by  the 
ratio  of  the  d-c  current  during  keying  to  the  current  with  key  closed  continuously.    If  the 
former  is  designated  IM  ,  and  the  latter  Ic,  the  mark-to-space  ratio  is 


LOW-POWER  TRANSMITTERS.  Modern  transmitters  having  a  power  output  of 
1  kw  or  less  are  usually  designed  as  completely  self-contained  units.  It  is,  accordingly, 
only  necessary  to  provide  power  line,  audio  input,  antenna  or  transmission  line,  and  any 
necessary  external  monitor  connections.  Such  transmitters  are  usually  designed  to 
operate  from  either  a  115-volt  or  a  230-volt  single-phase  a-c  source.  Power  and  audio 
circuits  may  be  conveniently  brought  into  the  transmitter  through  either  conduits  or 
trenches;  trenches  permit  somewhat  greater  flexibility  for  future  change  or  modification. 
External  wiring  and  switching  should  be  installed  in  accordance  with  fire  insurance  under- 
writers' specifications.  Ground  wires  should  be  connected  by  as  short  a  run  as  possible 
to  a  low-resistance  ground,  either  the  antenna  ground  system  if  any  portion  of  it  is  in- 
stalled near  the  transmitter  or  a  separate  ground  consisting  of  copper  plates  buried  in 
moist  soil.  Antenna  leads  or  transmission  lines  should  be  insulated,  not  only  for  the 
normal  operating  potentials  but  also  sufficiently  well  to  protect  against  induced  voltages 
caused  by  lightning  hits.  A  protective  gap  connected  between  antenna  and  ground  will 
discharge  heavy  induced  charges. 

HIGH-POWER  TRANSMITTERS.  High-power  transmitters  are  usually  designed  so 
that  the  equipment  is  segregated  in  accordance  with  a  functional  grouping,  so  that  the 
low-power  audio-frequency  equipment,  for  example,  is  well  isolated  from  the  high-power 
r-f  amplifiers.  Each  of  these  elements  has  somewhat  unique  installation  problems  and 
will  be  considered  separately. 

Audio  Equipment.  If  the  transmitter  is  located  some  distance  from  the  originating 
point  of  the  transmitted  intelligence  and  connected  to  that  point  by  telephone  line  or 
radio  relay,  it  is  necessary  to  provide  line  terminating  equipment  to  match  the  telephone 
line  properly  and  to  restore  the  signal,  attenuated  by  the  line,  to  its  original  amplitude. 
The  necessary  pads,  line  equalizers,  line  amplifier,  together  with  the  percentage-modula- 
tion meter  and  transmitter-frequency  monitor,  are  usually  mounted  in  a  standard  tele- 
phone-type rack  in  the  operating  room  where  they  may  be  conveniently  observed  by  the 
transmitter  operator.  It  is  not  ordinarily  necessary  to  provide  additional  external  shield- 
ing for  this  equipment;  however,  it  is  necessary  that  it  be  adequately  grounded  through  a 
low-impedance  and,  preferably,  separate  ground  lead.  The  incoming  audio-frequency 
circuits  and  the  power  supply  wiring  to  the  amplifiers  should  be  carefully  isolated  from 
each  other,  preferably  by  running  in  separate  conduits,  or,  if  in  a  common  trench,  by  run- 
ning the  audio  circuits  in  conduit.  Audio-frequency  circuits  should  use  twisted  pair 
enclosed  in  a  tight  external  shield. 

Low-power  Intermediate-radio-frequency  Amplifiers.  This  portion  of  the  equipment 
resembles  a  low-power  transmitter  and  requires  no  special  comment  beyond  the  points 
already  touched  upon  under  that  heading. 

High-power  Amplifier  and  Modulator.  The  installation  of  these  units  requires  careful 
planning  since  they  must  be  installed  so  that  they  may  be  conveniently  serviced  and  so 
that  the  required  high  voltages,  cooling  air  and/or  water,  and  r-f  output  connections  may 
be  conveniently  made.  Because  of  the  relative  simplicity  and  ease  of  maintenance,  the 
tread  in  design  of  high-power  amplifiers  has  been  toward  the  use  of  air  cooling  to  dissipate 
tfee  heat  generated  in  the  plate  circuits  of  the  r-f  amplifiers.  For  these  transmitters  provi- 


BIBLIOGRAPHY  7-137 

sion  must  be  made  for  an  intake  for  several  thousand  cubic  feet  of  air  per  minute,  and 
means  must  be  provided  to  exhaust  this  air  after  it  has  absorbed  the  heat  developed  in 
the  power  amplifier  tubes.  Filters  should  be  provided  either  in  the  transmitter  or  in  the 
air  inlet.  The  exhaust  air  is  frequently  piped  through  ducts  out  of  the  transmitter  into  a 
mixing  chamber  where  it  may  either  be  diverted  into  the  hot  air  heating  system  for  the 
building  or  exhausted  outside.  Because  of  the  large  volume  of  air  required,  the  inlet  duct 
is  frequently  installed  in  the  floor  below  the  transmitter.  It  is  possible  to  use  this  duct 
also  to  carry  interconnecting  wiring  out,  but,  because  of  the  danger  of  a  rapidly  spreading 
fire  in  case  of  an  accidental  short  circuit  in  this  wiring,  this  practice  is  not  advisable. 

Interconnecting  wiring  between  the  transmitter  units  should  preferably  be  carried  in 
separate  raceways  which  may  be  located  in  the  building  floor  or  arranged  as  troughs  on 
the  side  of  the  transmitter  cubicles.  All  such  raceways  should  be  arranged  so  that  they 
may  be  conveniently  opened  for  cleaning  purposes  and  to  permit  changes  in  wiring  as 
required. 

The  r-f  amplifier  should  be  provided  with  a  low-impedance  ground  connection  pref- 
erably in  the  form  of  a  wide  copper  strap  connected  directly  to  a  ground  consisting  either 
of  buried  metal  plates  or  a  connection  to  the  antenna  ground  system,  if  it  is  contiguous 
to  the  transmitter  building. 

Rectifier  and  Power  Equipment.  It  is  desirable  that  the  rectifier  tubes  be  located  so 
that  they  may  be  observed  by  the  operator  at  his  normal  position  during  the  operation  of 
the  equipment.  Rectifier  transformers,  however,  should  be  located  either  in  a  separate 
vault  or  in  a  transformer  yard,  enclosed  by  a  grill  or  fence  outside  the  transmitter  house. 
Fire  insurance  underwriters'  rules  should  be  consulted  and  closely  followed  since  these 
rules  differ  from  state  to  state  and  usually  cover  the  installation  of  this  type  of  equipment 
in  some  detail. 

SUBSTATION.  The  power  line  terminal  equipment  will  ordinarily  be  supplied  by 
the  power  company  having  contracted  to  supply  power.  This  equipment  will  include  any 
transformers  necessary  to  change  the  voltage  from  the  power  company  distribution  voltage 
to  the  voltage  required  at  the  transmitter  terminals,  together  with  necessary  circuit  dis- 
connects, circuit  breakers,  lightning  protectors,  and  metering  equipment.  To  reduce  the 
possibility  of  outages  resulting  from  a  failure  of  the  power  supply,  it  is  advisable,  if  pos- 
sible, to  provide  for  power  feeds  from  two  separate  sources.  Automatic  equipment  is 
available  which  will  switch  from  one  line  to  the  other  in  the  event  of  a  voltage  failure. 
To  secure  maximum  advantage  from  such  an  emergency  power  supply,  the  transmitter 
control  circuit  should  be  arranged  so  that  a  1-  or  2-sec  voltage  failure  will  not  necessitate  a 
complete  restarting  cycle.  Either  manual  or,  preferably,  automatic  reapplication  of 
plate  voltage  should  be  provided  after  a  no-voltage  drop  out,  provided,  for  most  rectifier 
tubes,  that  such  drop  out  does  not  exceed  2-sec  duration. 

BIBLIOGRAPHY 

Coleman  and  Trouant,  Recent  Developments  in  Radio  Transmitters,  RCA  Rev.,  Vol.  3,  316  (January 

•i  QOQA 

Doherty,  W.  H.,  A  New  High  Efficiency  Power  Amplifier  for  Modulated  Waves,  Proc.  I.R.E.,  Vol.  24, 

1163  (September  1936). 
Lee,  Reuben,  Radio  Telegraph  Keying  Transients,  Proc.  I.R.E.,  VoL  22,  213  (February  1934). 


SECTION  8 
FREQUENCY  MODULATION 


FREQUENCY-MODULATION  SYSTEMS 
ART<  BY  R.  D.  DUNCAN,  JR. 

1.  Fundamental  Relations 


PAGE 
02 


FRE  QUENCY-MODULATION 
TRANSMITTERS 
BY  J.  E.  YOUNG 

2.  Frequency-modulation  Broadcasting  ----     09 

3.  Frequency   Modulation  for  Emergency 

Transmitters  ......................     15 

4.  Transmitter  Circuits  .................     15 

FREQUENCY-MODULATION  RECEIVERS 
BY  LESLIE  F.  CURTIS 

5.  Comparison  with  Amplitude-modulation 

Receivers  .........................     16 


ART.  PAGE 

6.  Frequency  Detectors 19 

7.  Limiters 24 

DISTORTION  AND  INTERFERENCE  IN 

F-M  SYSTEMS 
BY  B.  D.  LOUGHLIN 

8.  F-m  Distortion  from  Non-uniform  Am- 

plitude and  Phase  Characteristics ....  26 

9.  Distortion  Due  to  Incomplete  Rejection 

of  Amplitude  Modulation 27 

10.  Distortion  Due  to  Multipath  Reception  29 

11.  Cross-talk  and  Beatnote  Interference.. .  29 

12.  Fluctuation  Noise  Interference 30 

13.  Impulse  Noise  Interference 31 


8-01 


FREQUENCY  MODULATION 
FREQUENCY-MODULATION  SYSTEMS 

By  R.  D.  Duncan,  Jr. 

In  frequency-modulation  systems  intelligence  is  communicated  by  variation  of  the 
frequency  or  phase  of  the  transmitted  wave  instead  of  its  strength  (as  is  done  in  ampli- 
tude modulation ).  Frequency  modulation  is  used  for  broadcasting,  for  fixed-tp-mobile 
stations,  and  for  one-  or  two-way  communication  in  services  such  as  police,  public-utility 
maintenance,  and  forest  patrol.  It  is  being  introduced  into  the  truck,  bus,  taxicab,  and 
railroad  fields.  So-called  "studio-transmitter  link'7  equipment,  which  provides  a  one- 
way radio  connection  in  lieu  of  telephone  lines  between  the  broadcast  studio  and  a  remote 
f-m  transmitter,  utilizes  frequency  modulation. 

During  the  war,  frequency  modulation  was  much  used  in  long-distance  relay  service, 
and  it  is  also  used  in  domestic  microwave  relay  service.  It  is  employed  in  the  allied  fields 
of  facsimile  and  television  broadcasting.  In  the  latter,  it  provides  the  sound  channel  and 
has  been  employed  for  relay  operation.  It  has  also  been  experimentally  used  for  power- 
line  carrier  current  communication. 

Frequency  modulation  is  not  well  adapted  to  circuits  subject  to  multipath  transmis- 
sion effects  since  serious  distortion  results  from  the  simultaneous  reception  of  several 
signals  differing  slightly  in  phase. 

1.  FUNDAMENTAL  RELATIONS 

The  modulating  signal  in  frequency  modulation  causes  the  carrier  frequency  to  be 
systematically  varied  above  and  below  the  unmodulated  value,  the  extent  of  the  varia- 
tion being  determined  by  the  strength  of  the  signal.  The  number  of  times  a  second  the 
frequency  is  so  varied  is  determined  by  the  frequency  of  the  signal. 

This  process  of  variation  of  the  carrier  frequency  produces  additional  frequency  com- 
ponents, called  sidebands,  which  lie  both  above  and  below  the  unmodulated  carrier  fre- 
quency. Theoretically,  there  are  an  infinite  number  of  such  upper  and  lower  sidebands 
which  differ  in  frequency  from  the  carrier  by  the  value  of  the  modulating  frequency  or 
frequencies  and  integral  multiples  thereof.  However,  then*  amplitudes  decrease  rapidly 
as  they  exceed  in  frequency  value  the  upper  and  lower  limits  of  the  maximum  frequency 
swing  imparted  to  the  carrier  so  that  satisfactory  reception  can  be  achieved  by  a  trans- 
mitting and  receiving  pass  band  somewhat  greater  than  twice  this  maximum  swing.  An 
important  characteristic  of  frequency  modulation  is  that  the  amplitude  of  the  carrier 
component  as  well  as  of  the  sideband  components  is  determined  by  both  the  amplitude 
of  the  modulating  signal  and  by  its  frequency,  or,  in  the  case  of  a  complex  wave  form,  by 
its  component  frequencies. 

Another  distinctive  feature  of  frequency  modulation  is  that,  theoretically,  the  power 
contained  in  the  carrier  and  the  infinite  number  of  sideband  components  is  a  constant 
value.  That  is,  the  transmitted  power  remains  unchanged  during  the  modulation  if  the 
circuits  are  such  as  to  transmit  all  the  sideband  frequencies.  Practically,  since  only  side- 
bands within  and  just  greater  than  the  maximum  frequency  swing  are  transmitted,  there 
will  be  slight  power  variation  during  modulation,  or  a  small  amount  of  amplitude  change 
or  modulation  will  accompany  frequency  modulation. 

SINGLE-FREQUENCY  MODULATION.  A  consideration  of  single-frequency  modula- 
tion will  serve  to  illustrate  the  essential  fundamental  characteristics  of  frequency  modula- 
tion and  the  difference  between  it  and  phase  modulation  (see  article  8-2  for  methods  of 
generating  frequency  and  phase  modulation). 

.  the  modulating  signal  is  a  single-frequency  component,  the  expression  for  a  f-m 
be  written  in  the  form 

E  cos  (<urf  -f  —  sin  pf)  (1) 

8-02 


FUNDAMENTAL  RELATIONS  8-03 

where  E  =  the  maximum  amplitude. 

a?  —  2  ir  times  the  unmodulated  carrier  frequency. 
p  —  2  TT  times  the  modulating  frequency. 
Aco  =  2  TT  times  the  peak  frequency  swing  of  the  carrier. 

The  corresponding  expression  for  a  p-m  voltage  is 

ep-m  —  E  cos  (cat  +  A0  sin  pt)  (2) 

where  A0  is  the  maximum  phase  or  angle  variation. 

The  difference  between  the  two  expressions  is  the  (sin  pt)  term.  For  frequency  modula- 
tion the  maximum  phase  or  angle  excursion  is  directly  proportional  to  the  peak  frequency 
swing,  i.e.,  to  the  strength  of  the  modulating  signal,  and  inversely  proportional  to  the 
value  of  the  modulating  frequency,  whereas  for  phase  modulation,  it  is  proportional  only 
to  the  strength  of  the  modulating  signal  and  is  independent  of  its  frequency. 

A  physical  conception  of  the  difference  between  frequency  and  phase  modulation  may 
be  had  from  Fig.  1.  This  shows  a  voltage  vector  E  which  without  modulation  during  an 
interval  of  time  t  has  rotated  counterclockwise  through  an  angle 
(o>i) .  If  it  is  assumed  that  an  observer  boards  the  vector,  as  it 
were,  so  as  to  rotate  with  it,  and  modulation  is  applied,  all  that 
will  be  noticed  is  a  rocking  back  and  forth  of  the  vector  through 
the  angle  (d=A0),  or  a  rotation  first  in  one  direction  and  then 
in  the  other  through  the  angle  ( =b A0) ,  which  may  greatly  ex- 
ceed 360  deg.  If  the  modulation  is  purely  frequency  type,  it 
will  be  noted  that,  with  a  fixed  signal  strength,  the  maximum 
angular  excursion  will  be  greater  for  the  low  modulating  fre- 
quencies than  for  the  high  ones.  If  modulation  is  of  the  phase 
type,  no  difference  in  the  maximum  excursion  of  the  vector 


will  be  noted  for  any  modulating  frequency.     If  the  modu-    FlG  l     Voltage  Vector  with 
lating  frequency  remains  fixed,  there  is  no  way  of  determin-          *  Phase  Modulation 
ing   whether   modulation  is  frequency  type  or  phase  type. 

If  amplitude  modulation  is  also  present,  a  slow  periodic  shortening  and  lengthening  of 
the  vector  would  be  noticed. 

In  the  discussion  on  the  mathematical  equivalent  of  discriminator  action  (p.  8-09)  it  is 
shown  that  the  mathematical  equivalent  of  f-m  detection  is  to  take  the  first  differential, 
with  respect  to  time,  of  the  variable  angle  through  which  the  vector  is  rotating.  For  eqs. 
(1)  and  (2)  this  angle  is  the  argument  of  the  cosine  function.  Doing  this,  the  two  follow- 
ing expressions  result. 

For  frequency  modulation 

—  (cot  -i sin  pt)  —  o>  4*  AOJ  cos  pt  (3) 

at 

For  phase  modulation 

—  (co£  -f  A0  sin  pt}  =  co  +  p&6  cos  pt  (4) 

The  recovered  signal  is  proportional  to  the  periodic  term  in  (3)  and  (4).  For  f-m  recep- 
tion, the  maximum  value  of  the  signal  is  proportional  to  the  extent  of  frequency  swing 
and  is  independent  of  the  value  of  the  modulating  frequency.  For  f-m  reception  of  a 
phase-modulated  wave,  the  maximum  value  of  the  signal  increases  directly  with  increas- 
ing modulating  frequency.  By  incorporating  frequency-distorting  circuits  at  the  trans- 
mitter or  at  the  receiver,  phase  modulation  may  be  converted  into  frequency  modulation, 
or  vice  versa. 

The  ratio  AOJ/J?  of  Eq.  (1)  for  frequency  modulation  was  originally  termed  the  "devia- 
tion ratio"  but  is  now  referred  to  as  the  "modulation  index,"  the  last  terminology  also 
applying  to  the  angle  A0  in  eq.  (2) . 

The  modulation  index  Aco/p  plays  an  important  part  in  the  theory  of  noise  suppression 
in  f-m  systems  and  involves  the  circuit  characteristics  of  both  the  transmitter  and  re- 
ceiver. The  FCC  has  established  ±75,000  cycles  (Aw  =  2?r  X  75,000)  as  100  per  cent 
modulation  for  an  f-m  broadcast  transmitter,  with  a  channel  band  width  of  200,000 
cycles.  It  requires  that  the  transmitter  be  capable  of  sustaining  this  maximum  peak 
frequency  swing  for  any  aural  modulating  frequency  between  50  and  15,000  cycles,  with- 
out exceeding  certain  specified  levels  of  harmonic  distortion. 

For  this  maximum  peak  frequency  swing,  the  modulation  index  Aco/p  would  have  a 
value  of  5  for  a  modulating  frequency  of  15,000  cycles  and  a  value  of  1500  for  50  cycles. 
There  would  be  five  upper  and  lower  sidebands  within  the  overall  swing  band  for  the 
higher  modulating  frequency  and  a  maximum  of  1500  sidebands  for  the  lower  frequency. 


8-04 


FREQTJENCY  MODULATION 


The  standard  for  100  per  cent  modulation  for  the  sound  channel  of  television  broad- 
casting is  ±25,000  cycles  with  the  same  aural  frequency  band  as  f-m  broadcasting.  It  is 
suggested  in  the  Standards,  however,  that  the  f-m  transmitter  be  designed  for  satisfactory 
operation  at  a  peak  swing  of  ±40,000  cycles.  In  the  case  of  point-to-point  communica- 
tion services,  such  as  police,  the  maximum  overall  swing  band  specified  by  the  FCC  is 
three-quarters  of  the  channel  band.  For  the  30-50  megacycle  band,  one  of  the  several 
allocated  for  these  services,  the  channel  width  is  40,000  cycles,  for  which  the  overall  swing 
band  would  be  30,000  cycles;  100  per  cent  modulation  would  then  be  ±15,000  cycles. 
Assuming  3000  cycles  as  the  maximum  modulating  frequency,  the  modulation  index  would 

Substituting  (P)  for  the  ratio  Aw/p  in  eq.  (1)  and  for  A0  in  eq.  (2),  the  equivalent  side- 
band form  of  expression  may  be  written  as  follows: 

e  »  E[JG(P)  cos  v*  -  Ji(P)  cos  (w  -  fit  +  Ji(P)  cos  (w  +  p)t 

-f  J2(P)  cos  (w  -  2p)£  +  ^s(P)  cos  («  -f  2p)t 

-  J3(P)  cos  (w  -  3p)£  +  Js(P)  cos  (w  -h  3p)i  (5) 
+  J4(P)  cos  (w  -  4p)«  4-  /4(P)  cos  (w  -r-  4p)t 

-  J6(P)  cos  («  -  5p)  *  4-  «/*(P)  cos  (co  +  5p)t 


+  J»(P){-  l)n  cos  («  -  np)t  +  Jn(P)  cos  (w  4-  np)i] 

The  coefficients  /.(P),  Ji(P),  J2(P)  -  -  •  J»(P)  are  Bessel  functions  of  the  first  kind,  of 
order  1,  2  •  •  •  n,  and  argument  (P).  Values  of  the  argument  (P)  for  frequency  modula- 
tion, as  is  shown  later  on,  may  vary  from  approximately  1  radian  to  the  order  of  1500 
radians.  For  phase  modulation,  the  maximum  value  for  broadcasting  is  5  radians. 

For  high  values  of  the  argument  (P) ,  the  approximate  value  of  a  particular  order  Bessel 
function  may  be  computed  from  the  expression  (see  also  Section  1). 

(6) 

Values  of  Bessel  functions  here  involved,  of  zero  order,  for  integral  values  of  the  argu- 
ment from  1  through  9,  and  for  orders  1  through  44,  corresponding  to  integral  values  of 
the  argument  of  1  through  29,  are  to  be  found  in  the  book,  Tables  of  Functions  by  Jahnke 
and  Emde.  Values  corresponding  to  decimal  values  of  the  argument,  increasing. by  incre- 
ments of  0.2  from  0.2  through  6.0,  and  in  increments  of  0.5  from  6.5  through  16.0  for 
orders  zero  through  13,  are  given  in  British  Association  for  the  Advancement  of  Science, 

Reports  on  the  State  of  Science,  1915, 

i*0ie— , — i — i — I 1 — i — : — 5 — i — i — i — i — i — i — i     on  The  Calculation    of  Mathematical 

Tables,  pp.  28-33. 

For  large  values  of  the  argument,  the 
reader  is  referred  to  Tables  of  Bessel 
Functions  Jn(x}  for  Large  Arguments 
by  M.  S.  Corrington  and  W.  Miehle, 
Journal  of  Mathematics  and  Physics 
(M.I.T.),  Vol.  24,  30  (February  1945). 
Argument  values  are  presented  in  vari- 
ous incremental  groupings,  extending 
from  29  through  300,  corresponding  to 
orders  zero  through  10.  Values  of  the 
specific  function  Jn(1000)  corresponding 
to  order  values  of  935  through  1035  in- 
creasing by  steps  of  1  are  given  in  Tables 
of  Bessel  Functions  J"«(1000)  by  M.  S. 
Corrington,  Journal  of  Mathematics  and 
Physics  (M.I.T.),  Vol.  24,  144  (No- 
vember, 1945). 

The  first  ten  orders  of  the   Bessel 
function  coefficients  of  eq.  (5)  are  plotted 
in  Fig.   2.     These   curves  show   that, 
(P)    or    modulation   indices,   Aco/p   or 
both    positive    and    negative   values 
of   any   sideband    component   and 


-0.4 


01234 


Y 


Z\ 


56     7    8    9    10  11  12  13  14  15  16 
Modulation  iadex  (P) 

(a) 

Fia  2a»     Modulation  Index  (P) 


depending   upon    the    value    of  the    argument 
M*  the    different    order    Bessel    functions    have 

and  therefore   pass   through  sero.    The   amplitude    ._ „ —^^^^   „,.., 

also  o€  the  carrier  may  be  zero,  that  is.  may  be  entirely  missing  from  the  f-m  or  p-m  wave! 
As  is  sfeown  later  on,  this  characteristic  as  it  relates  to  the  carrier  component  provides  a 
basis  for  measuring  the  extent  of  frequency  swing  or  the  degree  of  frequency  modulation. 


FUNDAMENTAL  KELATIONS 


8-05 


Further  illustrative  of  the  relative  amplitudes  of  the  carrier  and  sideband  components 
and  the  frequency  spread,  as  indicated  in  eq.  (5) ,  values  of  the  first  ten  order  Bessel  func- 
tions are  given  in  Table  1  for  six  values  of  the  argument  from  0.3  to  5.  It  is  observed  that 
for  values  of  the  modulation  index  of  0.5  and  less  the  amplitudes  of  the  sidebands  beyond 
the  first  do  not  exceed  approximately  3  per  cent  of  the  carrier.  For  a  modulation  index 


0.4 
0.3 
0.2 
0.1 
0 
-0.1 
-0.2 
-0.3 
-0.4( 

J3(P 

> 

-s 

^ 

(?)f 

Sl*tf 

/ 

/* 

\r 

~< 

\  / 

^ 

V*" 

-Je 

(P) 

f 

/ 

/ 

\ 

A 

\ 

\ 

-^ 

f 

/, 

/  , 

s\ 

i 

\ 

^ 

1 

s  , 

N" 

/^ 

s 

., 

•^ 

^ 

^ 

s 

\ 

\ 

\ 

\ 

1 

X 

/ 

k 

\ 

y 

\ 

> 

J 

\/ 

/ 

X 

\ 

^ 

\ 

AS 

A 

/ 

/ 

\ 

^ 

V 

^ 

^ 

^ 

*^- 

s 

"**^ 

)     1     23     4     5    6     78     9    10  11  12  13  14  15  1 
Modulation  index  (P) 

FIG.  2&.    Modulatioa  Index  (P) 


0.4 
0.3 


0.2 


-0.1 
-0.2 
-0.3 
-0.4, 


j'7(pK 

£e 

(Pj 

/-J 

5<p 

\ 

\ 

/ 

/ 

^ 

X 

JS 

N 

^ 

^ 

\ 

/ 

/ 

/, 

/^ 

/ 

\ 

\ 

\ 

\ 

/ 

^ 

s 

^ 

^ 

\ 

\ 

s 

S  , 

Y 

\ 

\ 

\ 

A 

/ 

^ 

y 

v,  / 

\j 

\ 

^v. 

^>. 

_»x 

~*s 

0    1     2     3    4    5     6     7    8     9   10  11  12  13  14  15  16 
Modulation  index  (P) 

(c) 

FIG.  2c.     Modulation  Index  (P) 

of  1.5,  the  third  order  sideband  is  approximately  12  per  cent,  and  the  fourth  order,  2  per 
cent,  of  the  carrier.  For  values  of  3  and  5  for  the  modulation  index,  all  sidebands  beyond 
the  sixth  and  eighth  respectively  are  less  than  1  per  cent  of  the  carrier  amplitude. 

Table  1.     Values  of  Bessel  Functions 


Order 


Argument  or  Modulation  Index  (P) 


71 

(P)  =  0.3 

(P)  -  0.5 

(P)  -  1.0 

(P)  -  1.5 

(P)  -  3.0 

(P)  =  5.0 

0 

-f  0.9770 

+  0.9380 

+  0.7650 

+  0.5050 

-0.26010 

-0.17760 

1 

4-0.1490 

+0.2410 

+  0.4401 

+  0.5600 

+  0.33910 

-0.32760 

2 

+  0.0112 

+  0.0300 

+  0.1149 

+  0.2330 

+  0.48610 

+0.04657 

3 

+  0.0006 

+  0.0025 

+  0.0195 

+  0.0615 

+0.30910 

+0,36480 

4 

+  0.0024 

+  0.0113 

+  0.13200 

+0.39120 

5 

+  0.04303 

+0.26110 

6 

+  0,01139 

+0.13100 

7 

+  0.00254 

+0,05338 

8 

+  0.01840 

9 

+0.00250 

Equation  (5)  with  Fig.  3  illustrates  additional  differences  between  frequency  and  phase 
modulation.  Figure  3  (a)  shows  a  carrier  and  sideband  components  for  a  modulation 
index  equal  to  5  radians,  and  for  a  modulating  frequency  of  15,000  cycles.  All  the  com- 
ponents in  this  and  the  accompanying  figures  are  plotted  as  positive  regardless  of  their 
polarity.  For  a  modulation  index  of  5  and  a  frequency  of  15,000  cycles,  the  peak  carrier 
frequency  swing  is  75,000  cycles.  Figures  3(&)  and  (c)  are  for  this  same  peak  frequency 
swing;  (6)  is  for  an  aural  frequency  of  10,000  cycles  and  a  modulation  index  of  7.5,  and 
(c)  an  aural  frequency  of  5000  cycles  and  modulation  index  of  15.  Figures  3(<2)  and  (e) 


8-06 


FREQUENCY  MODULATION 


are  for  a  fixed  value  of  the  modulation  index  of  5;  (d)  is  for  an  aural  frequency  of  10,000 
cycles  and  a  peak  frequency  swing  of  50,000  cycles,  and  (e)  for  a  5000-cycle  aural  frequency 
and  a  25,000-cycle  peak  frequency  swing.  Figure  3(a)  is  representative  of  both  frequency 
and  phase  modulation.  Figures  3(6)  and  (c)  are  representative  of  frequency  modulation 
and  Fig.  3(<Z)  and  (e)  of  phase  modulation.  . 

For  frequency  modulation  a  decrease  in  the  modulating  frequency  with  a  nxed  peak 
frequency  swing  increases  the  peak  angle  swing,  alters  the  respective  values  of  the  carrier 
and  sideband  components,  increases  the  number  of  important  sidebands  within  the  fre- 
quency swing  band,  and  bunches  the  sidebands  lying  just  outside  of  the  swing  band  closer 

•2TT  x  15,000 


0.4^W  = 

*>  «    X  /3jUwu;  \r  j  —  ii,  >/—*.<«  *  *«. 

0.3 

s 

0.2 
0.1 
0 

' 

{ 

0 

j     f 

O.3 

0.2 

0.1 

0 


V 
* 


FIG.  3.     Sideband  Distribution  for  FM  and  PM 

to  the  peak  swing  limits.  For  the  three  aural  frequencies  considered,  the  first  three  side- 
bands lying  just  beyond  the  swing  limits,  the  outer  one  of  which  is  less  than  5  per  cent 
in  value  of  the  unmodulated  carrier,  occupy  frequency  spaces  of  respectively  45,000, 
25,000  and  15,000  cycles. 

Going  to  an  extremely  low  modulating  frequency,  for  example,  75  cycles,  there  would 
be  a  maximum  of  1000  upper  and  lower  sidebands  within  the  swing  band  with  some 
appreciable  out-of-band  components  bunched  relatively  close  to  the  swing  limits.  Math- 
ematical analysis  of  this  particular  case  has  shown,  that  the  fifteenth  out-of-band  com- 
ponent which  would  be  displaced  by  1125  cycles  beyond  the  swing  limits  is  approximately 
1  per  cent  in  amplitude  of  that  of  the  unmodulated  carrier,  with  the  further  outlying  side- 
bands rapidly  approaching  zero. 

It  is  to  be  concluded  for  frequency  modulation  that  (a)  for  a  given  energy  content  the 
higher  modulating  frequencies  require  a  somewhat  greater  band-pass  range  from  a  circuit 
design  standpoint  and  (&)  with  decreasing  modulating  frequency  the  required  band-pass 
range  approaches  the  overall  frequency  swing  band  in  width.  This  last  explains  why  only 
the  actual  swing  band  need  be  considered  when  the  frequency  is  varied  at  a  very  slow 
ntfce,  for  example,  by  manual  change  of  the  capacitance  of  an  oscillator,  even  though  varia- 
tiGa  is  over  a  relatively  large  range.  It  also  explains  why  steady-state  analysis,  under  the 
assroopiion  of  a  sinusoidal  frequency  variation  at  a  low  rate,  may  be  employed  in  the 
study  of  f-m  circuits  yielding  results  which,  when  properly  interpreted,  are  sufficiently 
aeeairate  for  design  purposes  (see  article  2-8). 

F wfcheraioire  the  fact  that  in  the  usual  speech  or  music  programs  the  energy  of  these 
feigMreawnc^  tones  is  quite  low  keeps  the  modulation  low  and  hence  the  magnitude  of 
feaoaoaiies  <$  higher  order. 


FUNDAMENTAL  EELATIONS  8-07 

The  practical  result  of  these  factors  is  that  f-m  systems  work  very  well  with  the  pass 
band  only  slightly  greater  than  the  swing  band. 

In  the  case  of  phase  modulation,  decreasing  the  modulating  frequency,  with  a  fixed 
phase  swing,  decreases  the  overall  frequency  swing,  that  is,  bunches  all  sidebands  closer 
together,  but  alters  neither  the  number  of  sidebands,  their  respective  amplitudes,  nor  the 
amplitude  of  the  carrier. 

If  eq.  (5)  represents  an  f-m  voltage  wave,  the  rrns  value  is  given  by 

2J?-(P)  4-  2J22(P)  +  2JVKP)  4-  2/42(P)  4-  2/52(P)  +•••]**      (7) 

The  theory  of  Bessel  functions  shows  that  the  quantity  in  the  brackets  approaches  unity 
as  the  number  of  terms  becomes  large.  That  is,  the  rms  voltage  of  an  f-m  wave  or  the 
power  transmitted  remains  substantially  at  a  constant  value  regardless  of  the  extent  of 
modulation  if  a  sufficient  number  of  sidebands  are  present. 

TWO-FREQUENCY  MODULATION.  Corresponding  to  eq.  (1),  the  expression  for 
two-frequency  modulation  may  be  written  in  the  form 

et-m  =  E  cos  (cat  -i sin  pt  -{ sin  qt)  (8) 

P  Q. 

where  IS,  to,  p,  and  Aco  have  the  same  meanings  as  before  and  q  =  2w  times  the  second 
modulating  frequency.  The  equivalent  sideband  form  of  this  expression  is  given  by  eq. 
(9),  in  which,  as  before,  (P)  =  Aco/p  is  the  modulation  index  of  the  p/2ir  frequency,  and 
(Q) ,  substituted  for  Aw/tj,  is  the  modulation  index  of  the  g/2-Tr  frequency.  For  simplifica- 
tion, this  expansion  is  limited  to  the  third-order  Bessel  functions  of  the  two  arguments. 

The  result  of  two-frequency  modulation  is  the  production  of  major  sidebands  involving 
each  of  the  two  frequencies  and,  in  addition,  other  sidebands  involving  the  sums  and  dif- 
ferences of  the  two  frequencies  and  integral  multiples  of  the  same  in  different  combina- 
tions. The  amplitudes  of  all  sidebands  are  products  of  pairs  of  Bessel  functions.  Since, 
for  finite  values  of  the  argument,  the  values  of  all  Bessel  functions  are  less  than  unity,  the 
product  of  any  pair  will  be  less  than  the  individual  value  of  either  of  the  pair.  The  change 
from  single-  to  two-frequency  modulation  thus  greatly  increases  the  number  of  sideband 
components  and  the  frequency  spread  or  space  occupied.  However,  the  amplitudes  of 
all  sidebands  have  been  decreased  in  value  so  that,  while  the  frequency  spread  has  in- 
creased, the  outer  sidebands  have  very  low  amplitudes. 

A  similar  situation  holds  when  more  modulating  components  are  added.  That  is,  in 
going  from  a  single  sinusoidal  modulating  signal  to  one  of  complex  wave  form  the  number 
of  sideband  components  is  greatly  increased,  but  their  amplitudes  automatically  become 
readjusted,  or  the  total  modulation  is  in  effect  divided  between  the  components,  so  that 
the  overall  frequency  spread  for  all  practical  purposes  undergoes  little  or  no  change. 

e  =  J5yo(P)/o(Q)  cos  co£  -  Ji(P)Jr0(Q}{cos  (co  -  p)t  +  cos  (co  +  P)t] 

+  /2(P)/o(£)  {cos  (co  -  2p)«  4-  cos  (co  -f  2p)t} 
-  J3(P)/o(Q){cos  (co  -  3p)t  +  cos  (co  4-  3p)«} 


-  /o(P)/i(£){cos  (co  -  q)t  4-  cos  (co  4-  q)t} 
4-  JWVaCQMcos  (co  -  2g)t  4-  cos  (co  -f- 

-  </o(P)«/3«?){cos  (co  -  3q)t  +  cos  (co  +  3q)t} 


s  [«  -  (p  -  q}]t  +  cos  la  +  (P  - 
—  cos  [co  -  (p  +  tilt  -f  cos  [co  4-  (P  -f-  tf)#f 

cos  [co  -  (p  -  2q)]t  -  cos  fa  -f  <j>  - 
+  cos  [co  -  (p  +  2q)}t  -  cos  [u  4-  (p  + 
(P) /,(Q)  {cos  [co  -  (p  -  3<z)lt  +  cos  fc  4- 
-  cos  [co  -  (p  -f  32)1*  -  cos  |o>  -f-  (?  4- 


8-08  FBEQUENCY  MODULATION 


+  /2(P)/i(Q)  (cos  [to  -  (2p  -  q)]t  -  cos  [co  +  (2p  -  q)]t 
—  cos  [w  —  (2p  +  q)]t  —  cos  [w  -h  (2p  +  q)]t] 

4-  Ja(P)  ^(Q)!  cos  [w  —  (2p  —  2g)]i  -f-  cos  [w  +  (2P  — 
-|-  cos  [w  -  (2p  +  2<?)]i  +  cos  [a?  -h  (2p  +  2g)]i} 

-f  J2(P)Jz(Q)  {cos  [w  -  (2p  -  3g)]f  +  cos  [a?  +  (2?  — 
-  cos  [a>  -  (2p  +  3«)]«  4-  cos  [a>  4-  (2p  4-  3g)]i} 


-  J3(P)/!(Q)  {cos  [tf  -  (3p  -  q)]t  +  cos  [w  4-  (3p  -  Q)]t 

-  cos  [w  -  (3p  4-  «)]*  +  cos  [w  4-  (3p  +  ff)]*} 

-  Ji(P)/»(Q)  {cos  [a  -  (3p  -  2ff)]«  -  cos  [w  4-  (3p  -  2q)]t 

4-  cos  [w  -  (3p  4-  2<z)l«  -  cos  [a  4-  (3p  4-  2q)]t] 

-  Js(P)/s(Q){cos  I«  -  (3p  -  3g)]«  4-  cos  [w  4-  (3p  - 

-  cos  [a  -  (3p  4-  3fl)]«  -  cos  [w  4-  (3p  +  3g)]«| 


4-—]  (9) 

To  avoid  overmodulation  on  the  peaks,  the  average  modulation  with  a  complex  wave 
form  must  be  reduced.  The  necessity  for  such  a  reduction  is  well  known  in  broadcasting. 
MEASUREMENT  OF  FREQUENCY  SWING.  In  the  development,  testing,  and  prac- 
tical operation  of  f-m  equipment,  it  is  necessary  to  be  able  to  measure  the  extent  of  fre- 
quency or  phase  swing  or  modulation.  The  fact  that  the  zero-order  Bessel  function  J"o(P) 
passes  through  zero  with  increasing  values  of  the  modulation  index  as  shown  in  Fig.  2  is 
made  the  basis  for  one  useful  method  of  measuring  the  peak  frequency  or  phase  swing. 
In  Table  2  are  given  the  first  five  values  of  the  argument  (P)  or  modulation  index  for 
which  the  sero-order  Bessel  function  Jo(P)  becomes  zero.  Other  values  may  be  obtained 
from  the  references  previously  given. 

Table  2.     First  Five  Values  of  Argument  (P)  for  Which  J0(P)  =  0 

2.4048 

5.5201 

8.6537 
11.7915 
14.9309 

In  the  operation  of  this  method,  a  receiver  of  the  heterodyne  type  is  tuned  to  the  un- 
modulated carrier  so  that  a  beat  tone  of  several  hundred  cycles  is  obtained.  Single-fre- 
quency modulation  is  then  applied  to  the  transmitter,  the  frequency  being  of  a  much 
higher  order  than  that  of  the  beat  tone.  The  strength  of  the  modulating  signal  is  increased 
until  the  amplitude  of  the  beat  tone  drops  to  zero,  which  for  the  first  null  corresponds  to 
the  first  passage  through  zero  of  the  zero-order  Bessel  function  «/o(P).  From  Table  2,  the 
value  of  the  modulation  index  for  this  null  point  is  2.4048.  The  value  of  the  modulating 
frequency  (p/2x)  being  known,  the  peak  frequency  swing  (Aw/2ir)  is  obtained  from  the 
ratio  Aoj/p  =  2.404S.  Further  increase  in  strength  of  the  modulating  signal  will  develop 
additional  null  points  corresponding  to  which  values  of  the  modulation  index  are  given  in 
Table  2. 

Another  method  of  indirectly  measuring  the  frequency  swing  provides  a  visual  observa- 
tion on  the  screen  of  a  cathode-ray  oscilloscope,  of  the  carrier  and  sideband  components  in 
their  respective  amplitudes  and  positions  within  a  frequency  band  somewhat  greater  than 
twice  the  swing  band.  In  this  method,  the  f-m  signal  wave  is  heterodyned  to  an  inter- 
mediate frequency  of  2  Me,  amplified,  passed  through  an  a-m  detector,  and  supplied  to  the 
vertical  deflection  plates  of  a  cathode-ray  oscilloscope.  The  heterodyning  oscillator  is 
itself  frequency-modulated  by  means  of  a  reactance  tube,  with  a  25-cycle  linear-sweep 
signal  to  peak  values  of  plus  and  minus  100,000  cycles.  The  linear-sweep  voltage  is  also 
impressed  on  the  horizontal  deflection  plates  of  the  cathode-ray  oscilloscope. 

In  action,  the  f-m  signal  wave,  which  consists  of  its  carrier  and  sideband  components, 
la  m  effect  "scanned"  frequency-wise  from  100,000  cycles  below  to  100,000  cycles  above 
^be  csarrier,  at  a  rate  of  25  times  a  second.  When,  during  this  scanning  process,  a  side- 
band component  or  the  carrier  is  encountered,  a  voltage  pulse  is  impressed  on  the  verti- 
cal <l®Seetion  plates  of  the  oscilloscope.  As  the  linear-sweep  voltage  is  also  impressed  on 


FREQUENCY-MODULATION  BROADCASTING  8-09 

the  horizontal  plates,  the  particular  component  will  appear  in  its  proper  location  within 
the  overall  swing  band. 

THE  MATHEMATICAL  EQUIVALENT  OF  DISCRIMINATOR  ACTION.  In  theo- 
retical investigations  of  f-m  systems,  as  with  other  communication  systems,  the  end 
product  usually  sought  is  a  determination  of  the  form  of  the  received  signal  in  comparison 
with  that  of  the  signal  originating  at  the  transmitter. 

An  alternating  voltage  or  current  is  represented  by  a  vector  of  length  proportional  to 
the  maximum  or  rms  value,  revolving  at  an  angular  velocity  equal  to  2-n-  times  the  fre- 
quency, or,  in  customary  nomenclature,  <w  =  2irf.  In  a  given  time  t,  the  vector  will  have 
rotated  through  an  angle  of  (wi  =  2-rr  ft)  radians.  From  the  elementary  theory  of  the 
mechanics  of  planer  rotating  bodies,  it  is  known  that  the  angular  velocity  of  rotation  of 
any  point  in  the  body  about  the  axis  is  equal  to  the  first  differential  of  the  instantaneous 
angle  of  rotation  with  respect  to  the  time.  In  the  present  case,  this  would  be  d(ut)/dt  =  co 
=  2ir/;  that  is,  the  frequency  is  equal  to  the  first  differential  of  the  instantaneous  angle 
with  respect  to  time. 

In  conventional  a-c  theory,  a  vector  revolves  at  a  constant  velocity  as  the  frequency 
has  a  constant  value.  In  frequency  modulation,  the  angular  velocity  varies  with  time,  in 
accordance  with  the  modulation,  about  a  mean  or  unmodulated  value.  The  frequency 
obtained  by  differentiation  is  therefore  the  "instantaneous  frequency,"  which  is  what  is 
sought.  The  expression  for  the  instantaneous  frequency  may  be  a  simple  periodic  func- 
tion of  the  modulating  frequency  or  it  may  be  a  complex  function  requiring  algebraic  and 
trigonometric  operation  or  a  Fourier  analysis  to  break  it  down  into  the  fundamental  and 
harmonic  components. 

Expressions  are  given  above  for  a  single-  and  a  double-frequency-modulated  f-m  volt- 
age wave  and  single-frequency-modulated  p-m  voltage  wave.  These  are  as  follows: 


(Aw  \ 

art  -\  --  sin  pt  \ 

(Aco  Aw    .        \ 

o)t  -\  --  sin  pt  -\  --  sin  qt  ] 
P  9  / 


ep-m  —  E  cos  (coi  +  A0  sin  pt) 

The  first  differential  with  respect  to  the  time  of  the  arguments  of  the  three  cosine  func- 
tions are,  respectively,  (co  +  Aco  cos  pt)  and  (co  +  Aco  cos  pt  +  Aco  cos  qt)  and  («  -f 
j>A0cosp£).  With  linear  detection  the  demodulated  output  is  directly  proportional  to 
these  expressions.  With  square-law  detection,  the  demodulated  output  is  proportional 
to  the  squares  of  the  expressions  and  would  contain  second-harmonic  components  in  addi- 
tion to  the  fixed  and  fundamental  frequency  components.  With  a  balanced  discriminator 
and  differentially  connected  detectors,  both  the  fixed  and  second-harmonic  components 
cancel  out,  leaving  the  fundamental  frequency  components  of  double  amplitudes.  Linear 
detection  is  employed  almost  exclusively  in  commercial  practice. 

It  is  important  to  note  that  this  mathematical  method  of  recovering  an  f-m  signal 
implicitly  assumes  (a)  that  there  is  no  amplitude  modulation  and  (6)  a  linear  input-output 
characteristic  of  the  hypothetical  discriminator.  It  gives  only  the  relative  amplitudes  of 
the  frequency  components  but  tells  nothing  of  their  absolute  values.  If  amplitude  modu- 
lation is  present,  the  maximum  voltage  amplitude  E  in  the  preceding  expressions  is  -also 
a  function  of  the  time  and  must  be  taken  into  due  account.  Under  most  practical  condi- 
tions, it  may  be  assumed  that  the  effects  of  amplitude  modulation  are  removed,  by  some 
means  such  as  limiting,  leaving  only  the  variable  angle  to  be  investigated. 


FREQUENCY-MODULATION  TRANSMITTERS 

By  J.  E.  Young 

Transmitters  for  two  major  types  of  services  are  discussed  in  this  section.  These  are 
frequency-modulation  broadcasting,  which  includes  the  sound  transmitters  used  for  tele- 
vision broadcasting,  and  emergency  communication. 

2.  FREQUENCY-MODULATION  BROADCASTING 

Frequency-modulation  broadcasting  has  been  assigned  the  frequency  range  from  88  to 
108  me.  This  band  has  been  divided  into  100  contiguous  channels  with  carrier  frequencies 
starting  at  88.1  me  and  ending  at  107.9  me.  Transmission  in  each  channel  is  permitted 


8-10  FREQUENCY  MODULATION 

with  a  maximum  frequency  swing  of  75  kc,  which  is  designated  100  per  cent  modulation. 
To  insure  that  f-m  broadcasting  will  be  a  high-fidelity  service,  the  FCC  has  set  up  the 
following  standards  for  the  overall  transmitting  system  from  microphone  terminals  to 
antenna: 

1.  Distortion:  less  than  3.5  per  cent,  50  to  100  cycles,  2.5  per  cent,  100  to  7500  cycles 
at  25  per  cent,  50  per  cent,  and  100  per  cent  modulation;  and  3.0  per  cent,  7500  to  15,000 
cycles  at  100  per  cent  modulation. 

2.  Noise  level:  at  least  60  db  below  100  per  cent  modulation  in  the  band  50  to  15,000 
cycles. 

3.  Amplitude  noise:  at  least  50  db  below  100- per  cent  amplitude  modulation  in  the  band 
50  to  15,000  cycles. 

4.  Frequency  characteristic:  the  transmitting  system  shall  be  capable  of  transmitting 
a  band  of  frequencies  from  50  to  15,000  cycles.    Pre-emphasis  shall  be  employed  in  accord- 
ance with  the  impedance-frequency  characteristic  of  a  series  inductance-resistance  net- 
work having  a  time  constant  of  75  microseconds.    The  deviation  of  the  system  response 
from  the  standard  pre-emphasis  curve  shall  be  between  two  limits.    The  upper  of  these 
limits  shall  be  uniform  (no  deviation)  from  50  to  15,000  cycles.    The  lower  limit  shall  be 
uniform  from  100  to  7500  cycles,  and  3  db  below  the  upper  limit;  from  100  to  50  cycles, 
the  lower  limit  shall  fall  from  the  3-db  limit  at  a  uniform  rate  of  1  db  per  octave  (4  db  at 
50  cycles);  from  7500  to  15,000  cycles  the  lower  Emit  shall  fall  from  the  3-db  limit  at  a 
uniform  rate  of  2  db  per  octave  (5  db  at  15,000  cycles). 

Included  in  the  system  for  which  performance  is  thus  specified  are  the  microphone  pre- 
amplifier, mixers,  program  amplifier,  studio-to-transmitter  link,  which  may  be  wire  or 
radio,  transmitter  line  terminating  amplifier,  and  transmitter.  The  Radio  Manufac- 
turers Associatiom,  working  through  industry  committees,  has  specified  the  performance 
of  the  components  of  the  system  so  that  it  would  be  possible  to  combine  elements  of  dif- 
ferent manufacture  without  exceeding  the  distortion  or  noise  limits  specified.  For  the 
transmitter,  the  following  standards  have  been  agreed  to  in  the  industry: 

1.  Distortion:  audio  distortion,  including  all  harmonies  up  to  30  kc,  shall  not  exceed 
1.5  per  cent  rms  from  50  to  15,000  cycles,  and  shall  not  exceed  1  per  cent  rms  between 
100  and  7500  cycles.    Measurements  shall  be  made  at  25  per  cent,  50  per  cent,  and  100 
per  cent  modulation,  for  audio  frequencies  of  50,  100,  400,  1000,  and  5000  cycles,  also  at 
100  per  cent  modulation  for  audio  frequencies  of  7500,  10,000,  and  15,000  cycles. 

2.  Noise  level:  at  least  65  db  below  100  per  cent  modulation  in  the  band  50  to  15,000 
cycles. 

3.  Amplitude  noise:  at  least  50  db  below  100  per  cent  amplitude  modulation  in,  the  band 
50  to  15,000  cycles. 

4.  Frequency  characteristic:  shall  not  deviate  more  than  1  db  from  a  straight  line,  or, 
if  pre-emphasis  is  used,  from  a  75-microsecond  curve  from  50  to  15,000  cycles. 

F-M  TRANSMITTER — SCOPE.  Like  the  a-m  transmitter  covered  in  pp.  7-128,  7-137, 
the  f-m  transmitter  is  considered  to  consist  of  all  audio  equipment  operating  above  stand- 
ard telephone-practice  level  and  all  r-f  equipment  from  the  source  of  the  r-f  oscillation 
to  the  transmission-line  terminals.  Unlike  a-m  transmitters,  f-m  practice  is  to  modulate 
at  low  power  levels,  multiplying  the  frequency  and  power,  usually  many  times,  before 
reaching  the  transmitter  output. 

FREQUENCY  CONTROL  AND  MODULATION.  Frequency  control  and  modulation 
are  tied  together  since  modulation  is  accomplished  by  actually  swinging  the  frequency 
back  and  forth  in  accordance  with  the  modulating  signal.  To  accomplish  this,  and  still 
keep  the  center,  or  average,  frequency  within  tolerance  (±2000  cycles  for  f-m  broadcast- 
ing) is  one  of  the  chief  problems  of  f-m  transmitter  design. 

Methods  of  frequency  modulation  may  be  divided  into  two  basic  systems.  In  one  of 
these,  phase  modulation,  modulation  is  effected  at  some  point  in  the  circuit  following  the 
oscillator,  which  is  crystal-controlled.  Modulation  is  accomplished  by  changing  the 
phase  of  the  crystal-controlled  signal  at  a  rate  corresponding  to  the  desired  modulation. 
It  is  characteristic  of  p-m  systems  that  the  frequency  shift  is  proportional  to  the  modulat- 
ing frequency  aa  well  ass  to  its  amplitude.  To  convert  to  true  frequency  modulation  it  is 
necessary  to  compensate  the  frequency  characteristic  of  tbe  modulator,  therefore,  so  that 
the  amplitude  of  the  modulating  signal  is  inversely  proportional  to  its  frequency.  One 
system  of  phase  modulation  is  shown  in  the  block  diagram  in  Fig.  1.  A  crystal  oscillator 
liaving  a  frequency  of  approximately  200  kc  is  used  to  drive  a  balanced  modulator  in 
wfeich  the  modulating  signal  is  the  audio  frequency  to  be  transmitted,  and  whose  a-f 
cliaraeteristie  has  been  corrected  to  convert  from  phase  to  frequency  modulation  as 
described  alxmi.  Tfee  output  of  the  balanced  modulator  consists  of  two  f-m  currents 
whose  deviations  are  instantaneously  in  opposite  directions.  These  two  signals  are  multi- 
plied ia  frequency  81  times  through  two  separate  channels  of  multipliers.  One  of  the  re- 


FREQUENCY-MODULATION  BROADCASTING 


8-11 


sultant  signals  is  heterodyned  to  another  frequency  about  2  me  removed.  The  resultant 
frequency  and  the  output  of  the  second  multiplier  channel  are  then  recombined  and  their 
difference  recovered.  It  will  be  noted  that  the  difference  frequency  is  independent  of  the 
frequency  of  the  original  oscillator  and  depends  only  on  the  frequency  of  the  2-mc  het- 
erodyne oscillator.  Additional  multiplication  of  48  times  is  then  necessary  to  reach  the 
final  operating  frequency. 

The  phase  modulation,  at  the  point  of  its  introduction,  is  usually  restricted  to  less  than 
30°  in  order  not  to  exceed  permissible  transmitter  distortion.    In  the  method  described 


Multiplier 
(Six) 

16,200  kc 

Mixer 

+5 

rttfl 


FIG.  1. 


48 


0-=2  x  485=2  x  48  x  81/^—75  kc  max, 
/=transmitter  output  frequency. 
Frequency  Modulation  by  the  Phase-modulation  Method 


above,  since  two  outputs  are  obtained  from  the  balanced  modulator  having  phase  modu- 
lation in  the  opposing  sense,  a  total  phase  shift  of  60°  is  possible.  The  multiplication 
necessary  to  convert  this  phase  modulation  to  frequency  modulation,  having  a  frequency 
swing  of  ±75  kc,  was,  in  this  particular  example,  3888  times.  The  use  of  dual  p-m  chan- 
nels not  only  permits  doubling  the  frequency  swing  but  also  makes  it  possible  to  cancel 
out  the  effect  of  the  200-kc  crystal  in  determining  the  transmitter  frequency  stability.  In 
the  arrangement  shown,  the  carrier  frequency  stability  is  a  function  only  of  the  stability 
of  the  2-mc  oscillator.  Some  improvement  in  f-m  noise  level  is  also  obtained  at  the  same 
time. 

Other  methods  of  phase  modulation  have  been  developed  which  are  quasi-mechanical 
in  nature.  One  of  these,  for  which  a  special  vacuum  tube  has  been  developed  known  as 
the  "Phasitron"  tube,  is  used  in  some  commercial  f-m  transmitters.  The  block  diagram 
of  the  circuit  is  shown  in  Fig.  2.  A  crystal  oscillator  operating  at  approximately  230  kc 


Audio 


Crystal 
oscillator 

230  kc 

Network 

<Pl 

4>2            I 

1*to3* 

<M            \ 

FIG.  2.     Frequency  Modulation  Using  the  "Phasitron"  Tube 

(the  exact  frequency  depends  on  the  assigned  station  frequency)  is  coupled  to  a  network 
which,  by  means  of  tuned  circuits,  produces  three  outputs,  all  of  the  same  frequency  and 
separated  in  phase  by  120°.  These  outputs  are  applied  to  alternate  wires  of  the  deflec- 
tor grid  of  the  Phasitron  tube  in  such  a  manner  as  to  produce  a  rotating  field.  A  cathode 
and  two  anodes,  which  are  at  a  positive  d-c  potential  with  respect  to  the  cathode,  are 
arranged  so  that  electrons  are  drawn  from  the  cathode  and  focused  into  the  form  of  a 
tapered  thin-edged  disk.  This  disk,  with  the  cathode  for  its  axis,  lies  between  a  neutral 
plane  and  the  deflector  grid  structure  and  extends  out  to  anode  1.  The  three-phase  poten- 
tial applied  to  the  deflector  grid  structure  deflects  the  electron  beam  so  that  the  outer 
edge  of  the  electron  "disk,"  if  it  could  be  seen,  would  appear  to  have  a  sinusoidally  ser- 


8-12 


FREQUENCY  MODULATION 


rated  edge  which  rotates  about  the  cathode  as  a  center,  at  a  speed  determined  by  the 
three-phase  voltage  applied  to  the  grid.  Anode  1,  located  cylindrically  around  the  cathode, 
outside  the  periphery  of  the  deflector  grids,  has  24  evenly  spaced  Dholes  arranged  around 
its  circumference;  12  of  these  are  above  the  plane  of  the  electron  disk  and  12  below.  The 
rotating  serrated  edge  of  the  electron  disk  impinges  on  this  series  of  holes.  As  the  serra- 
tions move  around  the  periphery  of  the  electron  disk,  the  electrons  alternately  pass  through 
the  holes  to  anode  2  and  as  the  serrations  move  on  one-half  cycle  are  completely  blocked 
from  anode  2.  Thus  the  current  flowing  to  this  anode  varies  sinusoidally  at  the  crystal 
frequency.  It  follows,  therefore,  that  any  variation  in  the  angular  velocity  of  rotation 
of  the  electron  disk  will  result  in  a  phase  variation  of  the  output  current  from  anode  2. 

A  coil  is  placed  around  the  outside  of  the  tube  so  that  the  magnetic  field  resulting  from 
the  current  Sowing  in  the  coil  is  perpendicular  to  the  plane  of  the  electron  disk.  The 
electrons  traveling  radially  out  of  the  cathode  toward  the  anodes  through  this  field  have 
a  force  exerted  on  them  in  a  direction  perpendicular  to  their  path  and  perpendicular  to 
the  direction  of  the  magnetic  field.  Thus,  an  angular  displacement  is  introduced  in  the 
rotation  of  the  electron  disk  causing  phase  variation  in  the  output  of  anode  2  as  described 
above.  Consequently,  if  an  a-f  signal  current  of  the  proper  amplitude  flows  through  this 
coil  the  output  of  the  tube  will  be  phase-modulated  in  accordance  with  this  audio  fre- 
quency. When  there  is  no  audio  signal  input,  the  serrations  about  the  periphery  of  the 
electron  disk  will  rotate  at  a  constant  amplitude,  and  the  output  frequency  of  anode  2 
will  be  the  same  as  the  frequency  of  the  crystal. 

Phase  excursions  as  great  as  720°  are  possible,  in  this  system,  but  if  low  distortion  is  to 
be  achieved  the  phase  shift  must  be  considerably  restricted. 

DIRECT  FREQUENCY  MODULATION.  Direct  modulation  of  the  transmitter 
master  oscillator  may  be  accomplished  by  several  different  methods.  They  all  func- 
tion by  changing  the  reactance  of  the  frequency-controlling  part  of  the  oscillator  circuit 

_^   Fm 

output 


Modulator       —Screen  Oscillator 

FIG.  3.     Frequency  Modulation  by  Reactance  Tubes 

in  accordance  with  the  modulating  frequency.  One  of  the  commonest  of  such  systems 
is  the  use  of  a  tube  or  tubes  having  their  plates  connected  to  the  oscillator  circuit  and 
their  grids  excited  by  an  r-f  voltage  derived  from  the  oscillator  tank  and  shifted  in  phase 
90°  with  respect  to  the  oscillator  tank  voltage.  Plate  current  drawn  by  these  tubes  will 
then  be  90°  out  of  phase  with  the  oscillator  output  and  thus  has  the  characteristic  of  a 
positive  or  negative  reactance.  The  amplitude  of  this  plate  current  is  then  varied  by 
means  of  the  audio  signal,  which  is  also  impressed  on  the  grid,  and,  consequently,  the 
oscillator  frequency  shifts  in  accordance  with  the  a-f  signal.  The  basic  circuit  is  shown  in 
Fig.  Z. 

Another  method  of  achieving  frequency  shift  in  the  master  oscillator  is  to  connect  the 
grid  circuit  of  the  modulator  in  parallel  with  the  oscillator  tank.    See  Fig.  4.    In  this  cir- 


Modulator 
FIG.  4.     Freqtieacy  Modulation  by  Input  Capacitance  Variation 

curi^  frequency  modulation  results  from  the  fact  that  the  input  capacitance  of  the  modula- 
tor tube  is  a  function  of  its  plate  circuit  impedance  and  its  transconductance,  so  that 


FREQUENCY-MODULATION  BKOADCASTING 


8-13 


It  follows  that,  if  the  transconductan.ee  is  varied  by  the  audio  input  signal,  frequency 
modulation  in  accordance  with  the  audio  input  is  obtained. 

All  the  systems  of  direct  frequency  modulation  have  the  common  characteristic  that  the 
transmitter  frequency  control  is  effected  with  reference  to  a  separate,  highly  stable  oscilla- 
tor. These  control  systems  may  be  divided  into  two  categories,  those  in  which  the  restor- 
ing force  is  proportional  to  the  deviation,  and  those  in  which  full  restoring  force  is  de- 
veloped regardless  of  the  amount  of  deviation.  In  general,  the  latter  will  provide  the 
most  accurate  frequency  control  since  the  transmitter  output  frequency  will  be  an  exact 
multiple  of  the  reference  oscillator  frequency  if  the  system  is  functioning  properly.  The 
control  voltage  developed  by  the 
error  signal  in  either  system  may 
be  used  to  correct  the  master 
oscillator  frequency,  either  by 
changing  the  bias  of  the  tube  used 
to  effect  modulation  or  through 
an  electromechanical  device  driv- 
ing a  correcting  capacitor  in  the 
tank  circuit  of  the  oscillator. 

One  of  the  earliest  of  the  first 
type  of  systems  is  shown  in  Fig.  5. 
A  portion  of  the  output  of  the 
f-m  oscillator,  which  is  modu- 


Reactance 
modulator 

Modulated 
oscillator 

To 

ters 

^ 

^ 

Discrirr 

M 

Reference 
crystal 
oscillator 

FIG.  5.     The  Crosby  Method  of  Frequency  Control 


lated  by  reactance  tubes,  is  fed  into  a  mixer,  together  with  the  signal  from  the  reference 
crystal  oscillator.  The  beat  between  these  two  signals  is  fed  into  a  discriminator  tuned 
so  that  it  is  working  on  the  center  of  its  characteristic  when  the  frequency  of  the  modu- 
lated oscillator  is  the  exact  subrnultiple  of  the  transmitter  frequency.  As  the  modulated 
oscillator  drifts  away  from  the  correct  frequency,  the  beat  signal  fed  into  the  discriminator 
changes  frequency  accordingly  and  a  d-c  voltage  is  developed  by  the  discriminator.  This 
voltage  is  applied  to  the  grids  of  the  reactance  tubes  in  the  proper  sense  to  correct  the 
frequency  of  the  oscillator.  This  method  of  frequency  correction  may  be  thought  of  as  akin 
to  negative  feedback,  since  the  degree  of  correction  is  a  function  of  the  gain  through  the 
mixer  and  discriminator,  and  the  sensitivity  of  the  reactance  tubes. 

One  of  the  several  variations  of  the  second  type  of  system  is  shown  in  Fig.  6.  The 
master  oscillator  is  frequency-modulated  by  a  reactance  tube  or  other  electronic  means. 
A  sample  of  its  output  is  divided  in  frequency  by  a  locked-in  oscillator,  or  multivibrator, 
to  a  frequency  low  enough  so  that  the  carrier  frequency  will  not  vanish  for  any  percentage 
of  modulation  of  any  audio  frequency  in  the  normal  pass  band.  For  example,  in  one  trans- 
mitter the  modulated  oscillator  has  a  center  frequency  of  4  Me,  requiring  a  3-kc  swing  to 
achieve  a  75-kc  frequency  swing  at  the  transmitter  output  frequency.  The  center  fre- 
quency is  divided  by  256  so  that 
the  frequency  swing  reaches  a 
maximum  of  12  cycles  per  second, 
If  an  audio  modulating  frequency 
of  30  cycles  per  second  is  used, 
the  phase  shift  is  then  0.4  radian 
or  24°.  This  modulation  index 
reduces  the  carrier  to  0.96  times 
its  unmodulated  value;  conse- 
quently, a  carrier  frequency  com- 
ponent is  always  present  to  effect 
synchronization.  The  output  of 
the  reference  crystal  oscillator  is 
also  divided  in  frequency,  in 
order  to  use  a  crystal  which 
can  be  easily  manufactured,  and 
these  two  outputs  are  fed  into  a 


Reactance 
modulator 

Mod 
osc 

ulated 
Ilator 

>          -        To 

• 

10-cycIe 
low-pass 
filter 

Reference 
crystal 
oscillator 

Divider 

' 

Div 

der 

Balanced 
phase 
detector 

FIG.  6.     Phase  Detector  Method  of  Frequency  Control 


balanced  phase  detector.  If  the  two  inputs  to  the  phase  detector  are  exactly  90°  out  of 
phase  its  output  will  be  zero.  Any  shift  in  phase  difference  away  from  the  90°  point  will 
produce  an  output  voltage  with  a  positive  or  negative  sign  depending  on  whether  the 
phase  shift  is  greater  or  less  than  90°.  This  voltage  is  then  applied  to  the  modulator  tube 
grid  to  correct  the  frequency  of  the  master  oscillator. 

A  system  of  frequency  correction  utilizing  an  electromechanical  circuit  is  shown  in  block 
diagram  in  Fig.  7.  In  this  circuit  the  oscillator  is  modulated  by  reactance  tubes  in  the 
conventional  manner.  A  sample  of  the  output  of  the  modulated  oscillator  is  divided  by 
240  and  then  fed  into  the  grids  of  a  pair  of  balanced  modulators.  The  output  of  the  ref- 


8-14 


FREQUENCY  MODULATION 


erence  crystal  oscillator  is  divided  by  5  and  then  split  through  two  phase-shifting  net- 
works, arranged  to  provide  a  90°  phase  difference  between  the  two  branches.  One  of 
these  branches  is  fed  into  the  grid  circuit  of  one  of  the  balanced  modulators  and  the 
other  to  the  grid  circuit  of  the  second  balanced  modulator.  If  the  divided  down  signal 
from  the  f-m  oscillator  is  in  phase  with  the  divided  signal  from  the  reference  oscillator 
there  will  be  no  a-c  output  from  the  balanced  modulators.  However,  if  there  is  any  fre- 
quency difference,  an  a-c  voltage  will  be  developed  in  the  plate  circuits  of  each  of  the  bal- 
anced modulators  and  the  two  voltages  thus  obtained  will  differ  in  phase  by  90°.  These 


FIG.  7.    Electromechanical  Frequency  Control 

two  outputs  are  connected  to  the  two  windings  of  a  two-phase  motor,  and  the  shaft  of 
this  motor  is  arranged  to  drive  a  small  capacitor  connected  in  the  tank  circuit  of  the  f-m 
oscillator.  The  a-c  outputs  of  the  balanced  modulators,  resulting  from  a  frequency  devia- 
tion in  the  modulated  oscillator,  produce  a  rotating  field  hi  this  motor  which  tends  to  ro- 
tate thejvariable  capacitor  in  the  proper  direction  to  correct  for  the  frequency  error.  As 
soon  as  the  frequency  of  the  modulated  oscillator  has  been  restored  to  the  point  where  its 
submultiple  is  in  exact  synchronism  with  the  submultiple  of  the  crystal  oscillator,  the  a-c 
output  of  the  balanced  modulators  drops  to  zero  and  the  tuning  capacitor  comes  to  rest. 
Another  frequency-control  circuit  operating  on  a  different  principle  is  shown  in  the 
block  diagram.  Fig.  8.  In  this  circuit  the  outputs  of  the  master  oscillator  and  the  crystal 
oscillator  are  combined  in  a  pair  of  mixers,  in  one  of  which  the  input  from  the  crystal 
oscillator  differs  in  phase^by  9G°  from  the  input  to  the  other.  The  output  of  one  of  the 


1     Reactance    I         ^ 
modulalor    f 

Modulaied 

oscillator 

To 

multipliers 

f 

I     Intfift- 
f     rator 

Mixer- 

L 

•f45° 

phase 
shift 

amp 

r+->      *-*! 

Discnm-  I  {  DiscrLm-  1 
tnator    |   {     inator    t 

I       Refa 

r.ence 
stal 
ator 

Mixer- 
amp 

< 

-45° 
phase 
shift 

1      o^'i 

j     ~~~i  

1      Pulse     1 
1  generator  I 

FIG.  8.    Pulse  Counter  Frequency  Control 

mixers  is  fed  through  &  pulse  generator  and  thence  through  a  pair  of  discriminators.  The 
eferimiaators  are  biased  diodes.  The  bias  on  these  diodes  is  set  just  above  the  peak 
ralue  of  tte  output  of  the  second  mixer.  The  result  is  that,  when  the  pulses  add  to  the 
sine-wave  output  of  the  second  mixer,  the  bias  is  overcome  and  the  pulse  is  passed  through 
tins  diode.  Wfeea  the  pulse  subtracts  from  the  sine  wave  the  bias  prevents  the  diode  from 
©ondaefcing  and  the  pulse  is  not  passed.  This  arrangement  serves  to  separate  the  pulses 
into  two  ciixmits;  o»e  circuit  is  energised  by  one  pulse  for  each  cycle  of  beat  between  the 
master  oscilatoir  aod  the  crystal  oscillator  when  the  signal  frequency  is  high,  and  the  other 


TRANSMITTER  CIRCUITS 


8-15 


circuit  is  energized  by  one  pulse  for  each  cycle  of  beat  when  the  signal  frequency  is  low. 
The  outputs  of  the  two  discriminators 
are  fed  to  an  integrator,  which  is 
simply  a  large  capacitance  and  thence 
to  the  grid  circuit  of  a  cathode  follower. 
The  output  of  this  tube  is  connected 
to  the  tube  which  effects  frequency 
modulation,  and  thereby  controls  the 
frequency  of  the  master  oscillator. 
This  circuit  will  tend  to  hold  the  center 
frequency  so  that  the  frequency  swings 
higher  and  lower  than  the  correct  fre- 
quency by  the  same  total  number  of 
cycles.  The  correction  is  continuous, 
and  the  speed  at  which  a  frequency 
shift  is  reflected  in  a  correcting  voltage 
is  a  function  of  the  time  constant  of 
the  integrating  capacity. 

3.    FREQUENCY     MODULA- 
TION FOR  EMERGENCY 
TRANSMITTERS 

Phase  modulators,  corrected  to  ob- 
tain frequency  modulation,  are  used 
almost  exclusively  for  this  class  of 
service.  The  necessary  degree  of  phase 
modulation  can  be  achieved  with  rela- 
tively few  stages  of  multiplication  of 
the  f-m  signal.  This  results  from  the 
combination  of  two  requirements  that 
greatly  restrict  the  necessary  p-m 
angle.  The  first  of  these  is  that  a  com- 
paratively narrow  frequency  swing  is 
used,  varying  from  ±12.5  kc  in  the  25 
to  30  Me  band  to  =t22.5  kc  in  the  152 
to  162  Me  band.  The  second  is  that 
the  lowest  unattenuated  modulating 
frequency  need  be  no  lower  than  500 
cycles  per  second.  Thus,  if  a  phase 
swing  of  1  radian  can  be  obtained  in 
the  modulator,  a  multiplication  of  only 
45  is  necessary  to  achieve  a  frequency 
swing  of  22.5  kc.  Since  considerable 
distortion  is  tolerable  before  any  loss 
of  intelligibility  results,  phase  modu- 
lators having  much  more  inherent  dis- 
tortion may  be  used  for  this  class  of 
service  than  for  f-m  broadcasting. 
Overall  distortion  as  high  as  10  to  15 
per  cent  has  been  found  to  have  no 
effect  on  the  intelligibility  of  the  signal. 
Since  most  transmitters  of  this  type 
are  portable,  the  most  important  char- 
acteristics are  small  size  and  weight, 
and  the  minimum  number  of  tubes. 

4  TRANSMITTER  CIRCUITS 

Because  of  the  many  times  the  f-m 
signal  must  be  multiplied,  in  f-m 
transmitters,  to  obtain  the  necessary 
frequency  swing,  the  low  power  Fio.  9.  Coaxial  Tank  Circuit 


8-16  FREQUENCY  MODULATION 

stages  operate  through  the  frequency  range  covered  by  the  section  on  a-m  transmitters. 
Additional  care  is  required  in  f-m  transmitter  design  to  avoid  high-Q  circuits,  since  _  the 
band  -widths  required  are  greater  than  in  other  transmitters.  For  the  same  reason,  it  is 
desirable  to  use  circuits  that  have  symmetrical  phase-shift  characteristics  about  the  center 
frequency. 

Output  amplifiers  for  transmitters  of  power  above  250  watts  usually^  employ  tubes 
having  internal  capacitances  great  enough  to  necessitate  the  use  of  transmissi on-line-type 
tanks.  To  avoid  stray  fields  which  might  affect  the  operation  of  the  exciter  stages,  and  to 
simplify  the  problem  "of  keeping  the  transmitter  enclosure  and  outer  conductor  of  trans- 
mission lines  at  ground  potential,  these  circuits  are  preferably  made  in  the  form  of  con- 
centric lines.  One  such  typical  circuit  is  shown  in  Fig.  9.  In  this  illustration  a  triode  is 
used.  To  avoid  interaction  between  plate  and  grid  circuits,  the  grid  is  grounded  for  r-f 
voltages,  and  excitation  voltage  is  applied  between  ground  and  the  filament  of  the  tube. 
A  three-quarter  wave  tuned  transmission  line  is  used  and  is  in  turn  coupled  to  the  output 
of  the  driver  by  a  small,  single-turn  loop.  The  plate  circuit  is  a  coaxial  line,  in  which  the 
anode  of  the  tube  forms  a  continuation  of  the  inner  conductor.  The  line  is  tuned  by 
moving  a  by-pass  capacitor  provided  with  fingers  which  make  contact  with  the  inner  and 
outer  conductor,  along  the  line.  Output  coupling  is  obtained  by  positioning  a  loop  in  the 
space  between  the  inner  and  outer  conductors.  Control  of  the  tightness  of  the  coupling 
is  effected  by  changing  the  angle  of  the  loop.  Maximum  coupling  is  obtained  when  the 
loop  lies  along  a  radius  of  the  outer  conductor. 

BIBLIOGRAPHY 

Carson,  John  R,,  Notes  on  the  Theory  of  Modulation,  Proc.  I.R.E.,  Vol.  10,  57  (February  1922). 

Armstrong,  Edwin  H.,  A  Method  of  Reducing  Disturbances  in  Radio  Signaling  by  a  System  of  Fre- 
quency Modulation,  Proc.  I.RJB.,  Vol.  24,  6S9  (May  1936). 

Crosby,  Murray  G.t  Band  Width  and  Readability  in  Frequency  Modulation,  RCA  Rev.,  Vol.  5,  363 
(January  1941). 

Jaffe,  David  Lawrence,  A  Theoretical  and  Experimental  Investigation  of  Tuned  Circuit  Distortion  in 
Frequency  Modulation  Systems,  Proc.  I.R.E.,  Vol.  33,  318  (May  1945). 


FREQUENCY-MODULATION  RECEIVERS 

By  Leslie  F,  Curtis 

5.  COMPARISON  WITH  AMPLITUDE-MODULATION  RECEIVERS 

Conventional  f-m  receivers  use  the  superheterodyne  principle  and  differ  from  a-m  super- 
heterodynes mainly  in  the  i-f  amplifier  and  in  the  second  detector. 

The  intermediate  frequency  is  chosen  to  give  adequate  image  reduction  depending  on 
the  service.  An  intermediate  frequency  of  10.7  Me  is  suitable,  and  is  specified  as  an  RMA 
standard,  in  receivers  for  f-m  broadcasting  in  the  assigned  band  from  88  to  108  Me  since 
the  image  response  then  falls  outside  the  bands  having  assigned  services  which  are  liable 
to  interfere.  A  higher  intermediate  frequency  of  the  order  of  21.7  Me  is  required  for  the 
sound  channels  of  television  receivers  to  avoid  interference.  The  higher  frequency  is 
favored  in  receivers  incorporating  both  f-m  broadcast  and  television  facilities  since  the 
same  components  may  then  be  used  for  both. 

The  i-f  band  width  should  be  sufficient  to  pass  the  side  frequencies  at  the  maximum 
system  frequency  deviation  without  excessive  attenuation  of  the  power  in  any  of  them  to 
preserve  the  fidelity  of  the  modulation.  A  band  width  of  at  least  150  kc  for  a  reduction 
of  not  over  50  per  cent  on  the  overall  response  curve  at  maximum  deviation  is  required  in 


slightly  rounded  i-f  response  curve  is  preferred  since  double-peaked  curves  increase  the 
phase  distortion  within  the  receiver.  The  uniformity  of  response  over  the  required  band 
should  be  as  good  as  can  be  obtained  with  overcoupled  or  stagger-tuned  i-f  circuits  if  no 
limiter  is  used.  Receivers  for  special  purposes  which  utilize  narrow  system  deviations 
commonly  use  limiters,  and  operate  on  the  portion  of  the  i-f  response  curve  which  is  above 
50  per  cent  of  maximum. 

It  is  possible  to  reduce  the  frequency  deviation,  and  therefore  the  band  width  necessary 
for  tbe  i4  stages,  by  feeding  back  some  of  the  demodulated  audio  output  to  a  reactance 
tube  associated  with  the  superheterodyne  oscillator  to  cause  it  to  follow  partially  the 


COMPARISON  WITH  A-M  RECEIVERS  8-17 

original  frequency  deviation.    The  distortion  in  the  receiver  and  the  overall  noise-to-signal 
ratio  are  reduced  thereby,  although  the  system  is  rather  expensive. 

The  overall  gain  in  an  f-m  receiver  is  usually  greater  than  in  an  a-m  receiver  since  satis- 
factory reception  may  be  had  at  very  low  levels  of  input  to  the  antenna  terminals,  and 
since,  when  a  limiter  is  used,  the  input  voltage  at  its  grid  terminals  must  be  at  least  1  volt. 
In  general,  gain  is  uneconomical  in  the  r-f  stages  at  the  frequencies  assigned  for  f-m  broad- 
casting, and  practically  all  the  gain  is  usually  obtained  at  intermediate  frequencies.  The 
usual  i-f  plus  converter  gains  are  of  the  order  of  10,000.  The  band  width  required  lowers 
the  gain  per  stage,  and  usually  one  or  two  more  i-f  stages  are  required  in  an  f-m  receiver 
than  in  an  a-m  receiver  using  the  same  types  of  tubes. 

The  frequency  detector  directly  or  indirectly  converts  the  frequency  modulation  to 
amplitude  modulation  and  then  recovers  the  audio  signal  by  amplitude  detection  (see, 
however,  last  paragraph  of  article  8-7) .  Some  means  of  reducing  or  preventing  response 
to  spurious  amplitude  modulation  due  to  noise  and  due  to  the  variation  in  amplification 
as  the  frequency  is  deviated  over  the  i-f  pass  band  is  usually  associated  with  the  frequency 
detector.  This  may  be  an  amplitude  limiter  preceding  the  frequency  detector,  or  the 
frequency  detector  itself  may  be  of  a  type  which  is  non-responsive  to  amplitude  modu- 
lation. 

A  frequency  detector  with  a  balanced  output,  that  is,  one  in  which  the  net  rectified 
output  is  zero  at  the  mean  intermediate  frequency,  is  preferred  since  spurious  audio  output 
can  then  be  produced  only  during  deviation  of  the  frequency  due  to  the  desired  modula- 
tion and  is  masked  considerably  by  the  latter.  The  d-c  output  of  a  balanced  frequency 
detector  may  be  used  to  control  the  bias  of  a  reactance  tube  associated  with  the  oscillator 
and  thereby  furnish  automatic  frequency  control.  De-emphasis  circuits  to  compensate 
for  the  pre-emphasis  at  the  transmitter  (corresponding  to  the  voltage  across  an  inductance 
in  series  with  a  resistance  when  the  combination-  has  a  time  constant  of  75  microseconds) , 
and  tone-control  circuits,  are  usually  included  in  the  a-f  system. 

Antenna  input  systems,  the  superheterodyne  oscillator,  and  the  first  detector  are  usually 
the  same  in  f-m  receivers  as  in  a-m  receivers  for  about  the  same  transmitted  frequency 
except  that  there  is  more  tolerance  in  tuning  and  in  frequency  drift  in  f-m  broadcast 
receivers  than  in  narrow-band  receivers. 

Automatic  volume  control  may  be  incorporated  and  is  desirable  to  keep  the  voltage 
applied  to  the  input  of  the  limiter  or  non-amplitude-responsive  detector  at  a  level  which 
prevents  overall  response  to  rapid  variations  of  the  net  antenna  input  voltage  over  as 
wide  a  range  as  possible. 

Certain  types  of  f-m  broadcast  receivers  are  difficult  for  a  novice  to  tune  by  hand  since 
there  are  multiple  tuning  positions  where  there  is  almost  equal  volume  of  response  to  the 
desired  program.  Minimum  harmonic  distortion  is  obtained  in  only  the  position  which 
corresponds  to  the  most  linear  portion  of  the  frequency  detector  characteristic.  The  pro- 
gram is  demodulated  in  the  other  positions  by  the  slope  of  the  sides  of  the  i-f  response 
curve  and  sometimes  by  the  reverse  slope  of  the  skirts  of  the  frequency  detector  charac- 
teristic. A  greater  volume  of  even-harmonic  distortion  than  fundamental  often  is  pro- 
duced between  the  several  tuning  positions  for  maximum  response.  Accurate  tuning  is 
also  required  for  the  most  effective  reduction  of  response  to  impulse  noise. 

A  zero-center-indicating  meter  operated  by  the  d-c  output  of  a  balanced  frequency  de- 
tector makes  an  excellent  tuning  indicator.  It  indicates  zero  for  the  proper  tuning  posi- 
tion, and  the  direction  of  the  deflection  shows  the  direction  of  any  mistuning.  Twin 
electron-ray  tuning  indicators,  in  which  the  illuminated  portions  of  the  opposite  halves 
are  unequal  except  in  the  proper  position,  are  sometimes  used  in  broadcast  receivers. 

Receivers  for  operation  in  both  the  a-m  and  f-m  broadcast  bands  generally  use  many  of 
the  circuit  components  in  both  bands.  The  intermediate  frequency  for  the  f-m  section  is 
much  higher  than  for  the  a-m  section,  but  transformers  incorporating  tuned  circuits  for 
both  frequencies  are  quite  satisfactory.  The  r-f  and  converter  stages  tolerate  a  minimum 
of  switching  because  of  the  high  frequency  and  are  often  separate  for  the  two  bands  in  the 
more  expensive  receivers.  The  audio  amplifier  and  power-output  stages  are  usually  com- 
mon to  both  sectfons.  Particular  care  in  the  design  of  the  audio  amplifier  and  sound 
reproducer  is  justified  since  low  harmonic  distortion  and  excellent  signal-to-noise  ratio  are 
realizable  with  frequency  modulation. 

Figure  1  is  the  circuit  diagram  of  the  r-f,  i-f,  and  detector  portions  of  a  low-priced  fm-am 
receiver.  The  desired  band  is  selected  by  a  ganged  switch  for  the  r-f,  oscillator,  converter, 
avc,  and  detector  circuits.  Both  bands  are  tuned  with  a  two-gang  condenser  having  sep- 
arate stator  sections  for  the  two  ranges.  The  f-m  section  includes  a  broad-band  input 
transformer  for  a  300-ohm  transmission  line,  a  tuned  input  circuit  to  the  converter,  and 
delayed  avc  to  obtain  the  proper  input  level  for  the  ratio-type  frequency  detector.  The 
a-m  section  includes  a  condenser-tuned  low-impedance  loop  and  series  loading  coil,  means. 


8-18 


FREQUENCY  MODULATION 


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FREQUENCY  DETECTORS 


8-19 


for  additional  antenna  input  from  the  leads  of  the  300-ohm  line  in  parallel,  and  an  un- 
tuned input  stage  for  the  converter. 

Crosstalk  between  common-channel  f-m  stations  is  reduced  in  receivers  incorporating 
good  a-m  rejection  means  to  a  point  where  only  the  modulation  of  the  stronger  of  two 
stations  is  audible.  Cross-modulation  in  the  stages  of  an  f-m  receiver  due  to  the  non- 
linearity  of  tube  characteristics  is  relatively  small,  whereas  in  a-m  receivers  it  is  one  of  the 
major  causes  of  interference  at  certain  input  levels. 

Phase  modulation  differs  from  frequency  modulation  -only  in  the  manner  in  which  the 
frequency  modulation  index  (deviation  ratio)  is  caused  to  vary  with  the  modulating  fre- 
quency. In  phase  modulation  it  is  proportional  to  the  product  of  the  amplitude  and 
frequency  of  a  modulating  component,  whereas  in  frequency  modulation  it  is  propor- 
tional to  the  amplitude  of  the  component.  Receivers  for  phase  modulation  are  therefore 
like  receivers  for  frequency  modulation,  designed  with  an  i-f  band  width  suitable  for  the 
maximum  frequency  deviation  from  the  center  frequency,  but  with  an  audio  filter  follow- 
ing the  frequency  detector  to  restore  the  original  amplitude  of  the  modulating  com- 
ponent for  the  output.  Pre-emphasis  of  the  higher  audio  frequencies,  as  specified  by  the 
FCC  for  f-m  broadcasting,  gives  some  of  the  characteristics  of  phase  modulation  to  this 
portion  of  the  transmission. 


6.  FREQUENCY  DETECTORS 

Frequency  detectors  usually  consist  of  some  form  of  i-f  slope  filter  which  has  a  linear 
variation  in  output  voltage  with  frequency  deviation  to  the  maximum  assigned  deviation, 
preferably  fed  with  constant  current  from  the  last  i-f  amplifier,  and  followed  by  a  con- 
ventional amplitude  detector.  Detectors  employing  a  phase  shift  between  the  voltages 
applied  to  separate  grids  of  multielectrode  tubes  corresponding  to  the  frequency  devia- 
tion have  been  described  but  are  not  commonly  used.  Circuits  which  provide  frequency 
detection  in  a  single  tube  and  which  are  substantially  unresponsive  to  amplitude  modula- 
tion are  in  the  developmental  stage.  The  detectors  described  herein  have  had  commercial 
application. 

SIMPLE  SLOPE  FILTERS.  A  loosely  coupled  i-f  transformer  tuned  to  one  side  of  the 
mean  intermediate  frequency  and  operated  at  a  point  on  the  side  of  the  resonance  curve 
where  the  response  is  about  72  per  cent  of  maximum  provides  a  simple  slope  filter.  The 
conditions  differ  with  the  coupling,  but,  for  ex- 
ample, when  the  coefficient  of  coupling  between 
primary  and  secondary  windings  is  0.3/Q,  where 
Q  applies  to  both  primary  and  secondary  windings, 
the  variation  in  amplitude  is  substantially  linear 
over  the  range  in  frequency  in  which  the  amplitude 
varies  from  50  to  95  per  cent  of  the  maximum  for 
that  stage  alone.  The  frequency  characteristic  for 
this  condition  is  shown  in  Fig.  2.  The  Q  of  the  cir- 
cuits is  reduced  by  loading  so  that  Qf/F  is  0.2, 
where  /  is  the  expected  frequency  variation  from 
the  mean  and  F  is  the  mean  intermediate  fre- 
quency. The  frequency  for  maximum  response  for 
the  stage  is  separated  from  the  mean  frequency  by 
0.35  F/Q  and  may  be  either  above  or  below  it. 
Full  use  of  the  most  linear  portion  of  the  charac- 
teristic results  in  22.5  per  cent  amplitude  modula- 
tion, whereas  a  single  tuned  circuit  provides  only 
15  per  cent.  The  i-f  amplifier  is  tuned  to  flatness 
to  operate  about  one  of  the  operating  points  men- 
tioned. Final  demodulation  is  obtained  in  a  con- 
ventional diode  rectifier  following  the  filter. 


1.0 
0.9 
0.8 
0.7 
0-6 
0.5 
0.4 
0.3 
0.2 
0.1 
0 

c 

%, 

\ 

H- 

0.2- 

H 

H 

1 

K 

0.2 

N 

"1 

\ 

1 

\ 

N 

1 

^s 

\ 

^ 

X 

)    0.1  0.2  0.3  0.4  0.5 

0.6  0.7  O.S  0. 

FIG.  2.     Frequency  Characteristic  of  Slope 
Filter 


This  type  of  frequency  detector  is  not  often  used  since  it  provides  no  inherent  balance 
against  spurious  amplitude  modulation.  The  output  due  to  impulse  noise  in  this  type  of 
frequency  detector  is  much  more  disturbing  than  in  one  which  is  balanced  for  zero  output 
at  the  mean  intermediate  frequency. 

DISCRIMINATORS.  Discriminators  as  shown  in  Figs.  3(o)  or  3(&)  are  used  widely  as 
frequency  detectors  since  they  provide  an  inherent  balance  against  amplitude  modulation 
at  the  mean  intermediate  frequency  and  require  only  a  minimum  number  of  components 
and  adjustments.  The  i-f  voltages  applied  to  the  two  diodes  are  respectively  the  sum  and 
difference  of  the  primary  voltage  and  one-half  the  secondary  voltage  of  the  transformer, 


8-20 


FKEQUENCY  MODULATION 


T-f  Inpui 


Whioh  is  tuned  to  the  mean  intermediate  frequency.    Since  these  vcJtag* Differ  in  phase 
hv  00°  at  the  mean  frequency,  the  diode  voltages  are  then  equal.    The  rectified  voltages 

frequencies  since  the  secondary  voltage  then  lags  by  more  than  ^^rever^^tr^ 

quencies,  and  a  directional 
output  substantially  pro- 
portional to  the  instantane- 
ous frequency  deviation 
over  a  considerable  range  is 
obtained.  Thus  an  audio 
voltage  is  available  at  the 
output  terminals  which  has 
an  amplitude  proportional 
to  the  frequency  deviation 
of  the  applied  signal.  A  d-c 
output  which  depends  on 
the  difference  between  the 
mean  applied  frequency  and 
the  frequency  to  which  the 
unit  is  tuned  is  also  devel- 
oped if  these  are  unequal. 
Figure  4  illustrates  typi- 
cal shapes  of  characteristic 
curves  obtained  in  a  dis- 
criminator stage  when  the 
primary  and  secondary  in- 
ductances and  Q's  are  equal. 
The  curves  are  plotted  in 
terms  of  Qf/F,  where  f/F 
is  the  ratio  of  the  instan- 
taneous frequency  deviation 


-i-B 


FIG.  3.     Discriminator  Circuits 


t/ 


to  the  center  frequency.  The  actual  output  is  obtained  by  multiplying  the  relative  out- 
put by  VieQ-LcoI,  where  e  is  the  rectification  efficiency  of  the  diode,  L  is  the  inductance 
of  the  windings,  w  is  the  mean  intermediate  angular  frequency,  and  I  is  the  rms  value  of 
the  current  supplied  by  the  last  i-f  stage.  The  linearity  and  the  magnitude  of  the  output 
are  near  optimum  for  the  conditions  shown  when  the  product  of  the  Q  and  the  net  coeffi- 
cient of  coupling  between  windings  is  about  2.7  although  many  similar  curves  may  be 
obtained  when  the  primary  and  secondary 
inductances  and  Q's  are  unequal. 

It  is  important  in  obtaining  a  symmetri- 
cal characteristic  curve  that  the  secondary 
winding  and  circuit  be  symmetrical  with  re- 
spect to  ground.  This  requires  that  the  center 
tap  on  the  secondary  be  placed  properly,  that 
the  coupling  between  the  halves  of  the  sec- 
ondary be  close,  and  that  the  leads  to  the 
diodes  be  short  and  have  no  spurious  couplings 
to  other  parts  of  the  circuit.  The  diode  ca- 
pacitances should  be  equal  since  that  of  one 
aids,  while  that  of  the  other  opposes,  the  mag- 
netic coupling  between  the  windings.  The 
required  Q,  which  includes  the  effects  of 
diode  loading,  and  which  may  be  reduced  by 
circuit  loading  if  necessary  to  obtain  the 
proper  band  width,  may  be  estimated  from  the 
abscissas  of  Fig.  4  over  which  the  linearity 
is  satisfactory  for  the  expected  ratio  of  f/F. 
The  coupling  and  by-pass  condensers  C  are  made  only  large  enough  to  have  low  impedance 
at-  the  intermediate  frequency  while  their  impedance  at  audio  frequencies  is  large.  The 
caaarreat  through  each  diode  load  resistance  r  should  be  only  that  rectified  by  the  individual 
diode.  Any  shtmt  resistance  across  the  output  terminals,  such  as  ra,  is  made  large  so 
that  the  cwrent  through  it  and  the  two  resistors  r  in  series  will  not  bias  appreciably  the 
diode  delivering  the  smaller  instantaneous  output. 


0.5 

0.4 


0.3 


a  0.1 


-0.2 


-0.4 


1.0 


-(-1,0 


+  2.0 


FIG.  4.    Discriminator  Characteristics 


FREQUENCY  DETECTORS 


8-21 


The  i-f  choke  shown  in  Fig.  3 (a),  which  is  effectively  in  parallel  with  the  primary  in- 
ductance, may  be  omitted  if  the  diode  loads  are  connected  as  in  Fig.  3(&).  This  is  permis- 
sible if  the  desired  Q  can  be  obtained  with  the  increased  loading  effect  which  the  diodes 
present  for  this  connection.  Each  individual  diode  presents  a  load  of  r/1.8  in  Fig.  3  (a) 
and  r/2.8  in  Fig.  3(6),  where  r  is  as  indicated  in  the  figures,  and  the  diode  rectification 
efficiency  has  the  usual  value  of  about  90  per  cent.  The  desired  Q  may  be  obtained 
either  by  loading  the  windings,  as  with  r&  or  by  designing  the  individual  windings  with 
the  diameter  and  spacing  of  turns  which  result  in  the  required  value. 

The  frequency  at  which  the  response  is  zero  is  adjusted  by  tuning  the  secondary  circuit 
of  the  discriminator.  The  symmetry  of  the  response  curve  about  the  zero  point  is  ad- 
justed by  tuning  the  primary  circuit.  This  procedure  is  easier  than  adjusting  each  indi- 
vidual circuit  for  resonance  at  the  center  frequency. 

Some  special-purpose  receivers  are  arranged  to  respond  to  either  frequency  modulation 
or  amplitude  modulation  in  the  same  transmission  band  by  providing  a  reversing  switch 
for  one  of  the  diodes  of  the  discriminator.  "When  one  is  reversed  from  the  polarity  indi- 
cated in  Fig.  3,  the  sum,  rather  than 
the  difference,  of  the  rectified  voltages 
is  applied  to  the  output ,  which  is  proper 
for  the  demodulation  of  a-m  waves. 

SIDE-TUNED  CIRCUITS  AS  FRE- 
QUENCY DETECTORS.  Another 
frequency  detector  which  has  zero  out- 
put at  the  mean  intermediate  fre- 
quency consists  of  separate  tuned  cir- 
cuits C%Lv  and  CzLz  individually  tuned 
slightly  above  and  below  the  center 


4-B 


FIG.  5.     Frequency  Detector  with  Side-tuned  Circuits 


1.2 


,.1 


frequency  and  connected  as  shown  in 
Fig.  5.  The  response  curve  is  shown  in 
Fig.  6.  The  stage  is  tuned  by  adjusting  the  peaks  of  the  S  curve,  each  of  which  depends 
chiefly  on  the  tuning  of  one  of  these  circuits  but  is  influenced  slightly  by  the  tuning  of  the 
other.  An  alternative  tuning  method  is  to  transfer  the  connection  between  the  two  tuned 
circuits  from  the  full-line  to  the  dotted-line  connection  after  tuning  both  circuits  to  the 
center  frequency.  The  incremental  inductance  L  is  chosen  to  shift  the  resonance  by  the 
proper  amount  in  opposite  directions. 

Neglecting  the  coupling  between  the  two  side  circuits  caused  by  the  common  primary 
circuit  CiZ/i,  to  which  they  are  individually  coupled,  the  best -linearity  of  the  characteristic 

is  obtained  when  fi/F  —  0.75/Q,  where  A  is  the 
frequency  displacement  of  either  resonant  side 
circuit  from  the  center  frequency.  The  relative 
output  for  this  case  is  shown  in  Fig.  6  plotted 
against  relative  instantaneous  frequency  devia- 
tion Qf/F.  The  actual  output  depends  on  the 
coupling  to  circuit  CiZ/i.  The  capacitance  C\ 
of  the  driving  tube  may  be  tuned  out  at  the 
center  frequency,  and  the  effects  of  the  primary 
circuit  over  the  band  may  be  minimized  by 
loading  it  with  resistance.  The  presence  of  the 
tuned  primary  circuit  slightly  improves  the 
linearity  of  the  S  curve  near  its  peaks  by  an 
amount  not  shown  in  Fig.  6,  and  depending 
on  its  coupling  with  the  side  circuits.  To  ob- 
tain the  best  linearity  for  both  positive  and 
negative  deviations  it  may  be  necessary  to 
use  slightly  different  Q's  in  the  two  side  cir- 
cuits and  slightly  different  coefficients  of 


0.6 


-0.4 


-0.8 
-1.0 


7_ 


j_ 


^.O-O.a-o.6-0.4-0.2   0    0.2  0.4  0-6  0.8  i.o   coupling  between  the  primary  and  these  cir- 
***'  cuits.     Such  expedients   compensate  for  the 

Frequency  Characteristic  with  Side-  fact  that  the  resonant  peaks  of  the  side  circuits 
tuned.  Circuits  ,      ,   , ,  _. 

are  not  at  the  same  frequency. 

RATIO-TYPE  FREQUENCY  DETECTORS.  One  type  of  frequency  detector  which 
can  be  made  to  be  non-responsive  to  any  undesired  amplitude  modulation  has  been  called 
a  ratio  detector,  since  the  net  output  is  approximately  proportional  to  the  ratio  of  the  i-f 
voltages  applied  to  the  two  diodes,  although  the  process  by  which  this  is  accomplished  is 
indirect,  and  a  more  complete  analysis  shows  that  the  ratio  of  the  circuit  impedances 
rather  than  the  voltages  is  involved.  Two  of  many  possible  arrangements  are  illustrated 


8-22 


FKEQTJENCY  MODULATION 


I-f  input 


A.V.C. 


Ratio-type  Frequency  Detector  Circuits 


in  Figs.  7(o)  and  7(&).    In  Fig.  7  (a)  the  filter  is  quite  similar  to  the  discriminator  previously 
described  except  that  the  components  are  designed  to  tolerate  a  greater  a-c  load  current. 

In  Fig.  7(6)   the   side-tuned 
circuits  CzL®  and  CzL$  are  in 
^  series  with  no  magnetic  cou- 

p~1     J~^~      pling  to  the  choke  feed  L1? 
6H6/f~~K     "T'lcgi7"         while    capacitance    Cj   is   as 
-e* —       small  as  possible  to  reduce 
the    undesired    coupling    be- 
tween these  circuits.    In  both 
arrangements    condensers    C 
are  only  large  enough  to  have 
low  impedance  at  the  inter- 
mediate frequency. 

The  diodes  are  in  series  and 
are  therefore  forced  to  carry 
the  same  rectified  load  current 
which  charges  a  large  electro- 
lytic condenser  Ce  to  a  voltage 
which  depends  on  the  mean 
i-f  signal  level.  During  de- 
sired frequency  modulation  or 
un-desired  amplitude  modula- 
tion, this  voltage,  all  or  a  part 
of  which  may  be  used  also  for 
automatic  volume  control,  remains  constant  and  establishes  conditions  for  the  mean  rec- 
tification efficiency  of  the  diodes  and  for  a  voltage  drop 
in  the  a-c  impedance  of  the  source.  The  action  is 
illustrated  by  the  diagrams  of  Fig.  8  which  are  not  to 
scale  but  show,  diagrammatically,  superimposed  regu- 
lation curves  for  the  diodes  (one  inverted  to  show  the 
division  of  the  electrolytic  condenser  voltage  E}  for 
several  conditions.  Figure  8 (a)  shows  the  relations 
between  the  output  current  and  the  component  volt- 
ages which  lead  to  operation  at  a  particular  value  of 
outptit  current  at  one  input  level  and  frequency.  The 
peak  voltages  applied  to  the  input  terminals  of  the 
diodes  are  J£\  and  E$  respectively.  They  have  maxi- 
mum values  for  no  rectified  current  since  the  a-c  diode 
input  current  is  from  1.67  to  2.0  times  the  rectified 
current,  depending  on  the  rectification  efficiency  of  the 
diodes.  The  rectified  voltages*  whose  sum  is  Et  are  the 
input  voltages  multiplied  by  the  individual  rectification 
efficiencies  and  are  ei  and  &  respectively.  The  diodes 
are  in  series,  and  the  common-current  operating  point 
is  skown  by  the  marked  intersection. 

The  a-c  diode  input  current  is  in  phase  with  the  in- 
put voltage,  but  the  filter  networks  carry  substantial 
reactive  components  of  current.  Furthermore,  the  re- 
lations between  the  diode  input  and  output  currents  and 
voltages  are  non-linear.  Therefore  the  characteristics 
cautnot  be  expressed  readily  in  terms  of  the  circuit  con- 
stants. However,  each  circuit  and  associated  diode  has 
a  definite  regulation  curve  for  each  frequency  and  for 
each  current  level  supplied  by  the  last  i-f  tube.  The 
effective  impedance  whidb  determines  the  output  volt- 
age drop  in  terms  of  the  output  current  depends  pri- 
marily on  the  impedance  of  the  tuned  circuits  and 
secondarily  on  nonlinear  functions  of  the  current  and  Rectified  output  current 


FIG.  8.     Regulation  Curves  in  Ratio- 
type  Frequency  Detectors 


Rectified  output  current 

(a) 


Rectified  output  current 

(&) 


8(&)  stows  the  effect  of  deviating  the  fre- 

Tfefe  changes  the  filter  impedances  and  con- 
sequently  the  effective  output  impedance  and  the  slope 
«i  tin*  insulation  curves.    The  difference  between  either  diode  output  voltage  and  its  mean 
vafafe,  M/Z,  vsunes  ai  th«  modulating  frequency  and  represents  the  desired  audio  output. 


FREQUENCY  DETECTORS 


8-23 


Figure  8(c)  shows  the  effect  of  amplitude  modulation  of  the  input  signal  at  a  given 
frequency.  The  voltage  E  remains  constant  at  its  mean  value  by  virtue  of  the  charge  on 
the  electrolytic  condenser.  The  output  current  varies  instantaneously  with  amplitude 
modulation  although  its  mean  value  is  substantially  constant.  If  the  degree  of  inward 
modulation  is  sufficient  to  reduce  the  rectified  current  to  zero  as  illustrated  by  the  dotted 
lines,  rectifying  action  is  lost  and  intolerable  distortion  results.  The  resistance  r  is  made 
smaller  than  in  conventional  discriminators  to  allow  for  a  margin  in  inward  modulation 
without  reaching  these  limits.  The  ratio  of  the  uniform  signal  to  the  instantaneous  signal 
which  will  produce  cut-off  is  the  overdrive  and  should  be  at  least  2  to  1.  It  is  particularly 
important  that  the  relations  be  so  chosen  that  the  detector  is  not  cut  off  when  the  instan- 
taneous voltage  swings  to  its  lowest  value  as  the  frequency  is  deviated  over  the  selectivity 
response  curve  of  the  receiver. 

It  can  be  shown  that  with  ideal  diodes  with  rectification  efficiencies  of  100  per  cent,  and 
with  non-reactive  source  impedances,  the  output  voltage  e  is  independent  of  amplitude 
modulation  when  the  short-circuit  currents  I8,  as  shown  by  the  regulation  curves,  are  the 
same  for  the  two  diodes.  The  output  is  then  e  =  Q.5E(Zi  —  Z£/(Zi  +  Z2),  where  Z\ 
and  #2  are  the  impedances  of  the  two  filter  sections.  In  actual  ratio-type  frequency  de- 
tectors the  phase  angle  of  the  filter  impedance 
varies  with  the  frequency  deviation  and  the 
diode  efficiency  depends  on  both  applied 
voltage  and  rectified  current.  It  is  therefore 
difficult  to  compensate  perfectly  for  ampli- 
tude modulation  over  a  wide  range  of  either 
signal  level  or  deviation.  The  best  combina- 
tion of  impedances  is  determined  by  trial 
.For  the  circuit  of  Fig.  7(a),  for  example,  opti- 
mum values  of  coil  inductance  L%  and  L\, 
mutual  inductance  m,  resistance  r,  and  capac- 
itance C  will  be  found  for  a  particular  type 
of  tube  and  range  of  operating  levels  for  the 
best  linearity  of  desired  output  and  reduction 
of  amplitude  modulation. 

Some  of  the  possible  detector  characteris- 
tics for  simultaneous  amplitude  and  frequency 
modulation  are  illustrated  in  Fig.  9.  The 
curves  drawn  with  heavy  lines  show  the  output 
during  maximum  outward  amplitude  modu- 
lation. Figure  9  (a)  is  for  a  conventional  dis- 


FIG.  9.    Amplitude  Compensation  in  Ratio-type 
Frequency  Detectors 


criminator  without  the  compensating  effect  of  the  ratio-type  detector.  Figures  9(6)  and 
9(c)  are  for  partial  and  overcompensation  respectively.  In  (c)  high  input  level  produces 
less  output  over  a  portion  of  the  deviation  range  than  lower  input. 

The  diodes  carry  components  of  current  at  the  second-harmonic  frequency  as  well  as 
at  the  fundamental  frequency.  This  current  returns  through  the  filter  sections  and  pro- 
duces a  small  second-harmonic  voltage  which  shifts  the  effective  phase  of  the  peak  voltage 
to  be  rectified.  This  effectively  detunes  the  filter  sections  synchronously  with  the  ampli- 
tude modulation  and  accounts  for  an  unbalanced  characteristic  as  illustrated  in  Fig.  9(c£) 
between  two  signal  levels.  When  the  variation  in  input  level  is  due  to  deviating  over  a 
non-uniform  selectivity  response  curve  of  the  receiver,  the  resulting  characteristic  may  be 
as  shown  by  the  dotted  line. 

The  demodulated  signal  rises  and  falls  with  the  applied  signal  when  it  is  varied  slowly, 
as  in  tuning.  This  is  an  advantage,  since  the  proper  tuning  position  is  then  indicated  by 
maximum  volume. 

Since  a  large  degree  of  reduction  of  amplitude  modulation  is  obtained  in  the  ratio-type 
detector  stage  itself,  limiting  in  the  previous  stages  is  not  always  required,  and  the  signal 
level  at  the  last  i-f  tube  need  not  be  as  high  as  in  receivers  using  limiters.  Automatic 
volume  control  may  provide  sufficient  control  of  signal  level. 

Multipath  transmission  through  space  of  the  signal  applied  to  the  antenna  terminals  of 
the  receiver  may  result  in  amplitude  modulation  sufficient  to  reduce  the  instantaneous 
input  to  the  diodes  in  a  ratio-type  detector  below  their  cut-off  level,  and  in  this  case  a 
receiver  having  high  i-f  gain  followed  by  limiter  is  superior. 


8-24 


FREQUENCY  MODULATION 


r+B 


FIG.  10.     Grid-bias  Limiter 


7.  LIMITERS 

An  ideal  limiter  or  limiting  system  for  operation  at  the  intermediate  frequency  of  an 
f-m  receiver  delivers  an  rms  output  which  is  independent  of  the ^input  when  the  input  is 
above  a  threshold  level.  (However,  a  proposed  type  "dynamic  limiter  gives  output 
proportional  to  the  average  input  but  wipes  off  any  a-f  amplitude  modulation.)  It  should 
operate  instantaneously  and  therefore  should  not  include  time-constant  circuits  which 
delav  its  recovery  after  being  subjected  to  a  high  input  voltage,  as,  f  or  ^example,  a  burst 
of  impulse  noise.  The  loading  effect  of  the  limiter  on  associated  tuned  circuits  should  not 

change  with  signal  level.  ,  -,  i     •  i_  -xi_ 

The  greatest  value  of  a  limiter  is  in  reducing  amplitude  modulation,  synchronous  with 
the  desired  frequency  modulation,  which  may  be  introduced  by  the  deviation  of  the  fre- 
quency over  symmetrical  but  slightly  rounded  i-f  response  curves.  It  is  also  of  value  in 
reducing  random  or  impulse  noise  which  occurs  while  the  carrier  is  deviated  from  the 

GRID-BIAS  LIMITERS.  Limiters  in  which  the  operation  is  controlled  by  the  bias 
developed  by  grid  rectification  are  more  commonly  used  than  other  types.  The  funda- 
mental component  of  the  plate  cur- 
rent of  a  grid-bias  limiter  is  main- 
tained substantially  constant  over  a 
considerable  range  of  input  (usually 
about  10  to  1)  by  a  proper  correla- 
tion of  the  bias  developed  at  these 
levels  with  the  bias  necessary  to  cut 
off  the  instantaneous  plate  current,  . 
thereby  compensating  for  the  change 
in  angle  in  each  cycle  over  which  the 
plate  circuit  is  conducting.  This  re- 
lation is  most  easily  obtained  in  a 
pentode  tube  since  its  plate  current 
is  nearly  independent  of  the  plate  load.  A  typical  circuit  is  shown  in  Fig.  10.  The  out- 
put is  applied  to  a  circuit  tuned  to  the  fundamental  frequency  (usually  the  discriminator) , 
and  the  harmonics  are  filtered  out.  A  sharp-cutoff  tube  is  used  since  limiting  may  then 
be  obtained  at  low  input  levels. 

A  typical  grid-bias  limiter  static  characteristic  is  shown  by  the  solid  line  in  Fig.  11.  If 
the  grid  resistor  is  too  small,  the  efficiency  of  grid  rectification  and  the  developed  bias  are 
too  small,  so  that  the  angle  of  plate  conduction  is  not  reduced  sufficiently  at  high  input 
levels,  and  the  output  current  rises,  as  shown  by  the  upper  curve.  Conversely,  if  the  grid 
resistor  is  too  large,  the  output  falls  as  shown  by  the  lower  curve.  At  very  high  input 
levels  the  proper  relations  cannot  be  held  for  any  proportions  and  the  output  again  rises. 
Although  the  static  characteris- 
tic of  a  grid-bias  limiter  may  be 
made  flat  over  a  wider  range 
than  shown  by  placing  resistors 
by-passed  to  ground  in  the  plate- 
or  screen-supply  circuits,  the 
overall  operation  in  the  presence 
of  impulse  noise  is  not  satisfac- 
tory since  the  conditions  follow- 
ing a  burst  of  noise  are  not 
normal  and  the  output  suffers 
during  the  recovery  time  of  the 
plate  or  screen  circuits. 

An  approximate  rule  for  flat 
limiting  which  holds  for  pentode 
tubes  may  be  used  if  the  co- 
efficient a  in  the  expression  %  =  ae$&  for  the  grid  current  ig  in  terms  of  the  instantaneous 
applied  grid  voltage  e  is  known.  The  product  arJG?eH  should  be  35  or  40,  where  r  is  the 
grid  resistor  in  ohms  and  Ec  is  the  d-c  grid  voltage  necessary  to  cut  off  the  plate  current 
with  the  screen  voltage  selected.  The  rms  plate  current  is  then  approximately  Q.5gmEc 
in  the  range  of  rms  input  voltage  from  0.7  to  7.01<?c,  where  gm  is  the  transconductance  of 
the  tube  with  small  negative  bias  at  the  screen  voltage  selected.  The  level  at  which  limit- 
ing occurs  with  common  tubes  is  ordinarily  between  1  and  3  volts  but  may  be  adjusted 
over  a  small  range  by  selecting  the  screen  voltage. 


~o 

-o 

c 

«    o 
if 

^1 

Y 

,-<: 

^ 



~~~           -—  * 

**^ 

_..  «-»•"'"" 

Relailve  outout  vol 

s 

- 

"""""--— 

—  

/_ 

0          -10              0           -t-10           +20           +30          +4C 
R&lative  input  voltage  In  decibels 
FIG.  11.    Grid-bias  Limiter  Characteristics 

BIBLIOGRAPHY  8-25 

Cascaded  limiters  are  often  used  to  cover  a  greater  range  of  input  signals  over  which 
limiting  is  effective.  The  voltage  applied  to  the  grid  of  the  last  limiter  from  that  developed 
in  the  tuned  output  circuit  of  the  previous  limiter  is  made  to  fall  at  a  point  below  the  final 
upward  curvature  of  the  static  characteristic  so  that  an  additional  10-to-l  range  in  level 
may  be  handled.  All  the  i-f  tubes  may  act  as  limiters  when  the  i-f  amplifier  is  used  only 
for  f-m  signals.  The  rectified  d-c  voltage  in  the  grid  resistor  may  be  filtered  and  used  for 
avc  bias. 

The  time  constant  of  the  r-C  combination  in  the  input  circuit  of  a  grid-bias  limiter  should 
be  as  short  as  is  consistent  with  little  loss  of  i-f  gain.  The  recovery  time  is  then  fast  enough 
so  that  the  program  is  not  eliminated  for  an  audible  interval  after  a  burst  of  impulse  noise. 

PLATE  LIMITER.  A  plate  limiter  operates  when  the  plate  voltage  of  a  triode  or 
pentode  swings  downward  to  that  portion  of  the  plate-current  plate-voltage  characteristic 
where  an  increase  in  grid  voltage  in  the  positive  direction  produces  no  increase  in  plate 
current.  The  stage  is  operated  with  very  low  plate-supply  voltage  and  with  a  high- 
impedance  plate  load.  During  negative  grid  voltage  swings  the  plate  current  is  cut  off. 
The  instantaneous  output  current  swings  between  the  maximum  and  zero,  and  tends  to 
deliver  a  rectangular  wave  at  high  input  levels.  The  rms  output  voltage  increases  slightly 
with  increase  of  input  level  until  the  output  current  has  assumed  the  rectangular  wave 
form,  and  is  then  somewhat  less  than  half  the  plate-supply  voltage.  The  harmonics  of 
voltage  in  the  output  are  eliminated  in  the  tuned  circuit. 

Heavy  grid  current  loads  the  input  circuit  of  a  plate  limiter  during  positive  grid  swings. 
This  may  be  restricted  by  a  resistor  in  series  with  the  grid  lead  but  may  still  influence  the 
selectivity  and  gain  of  the  input  tuned  circuit.  A  plate  limiter  has  the  advantage  of  rapid 
recovery  time  if  the  plate-  and  screen-supply  voltages  are  not  influenced  by  the  tube  load 
but  is  more  often  used  in  clipping  and  shaping  pulses  than  in  amplitude  limiting  in  f-m 
receivers. 

LOCKED-IN  OSCILLATOR.  An  oscillator  operating  at  the  intermediate  frequency 
or  some  submultiple  thereof  may  be  used  to  cause  the  receiver  to  be  non-responsive  to 
amplitude  modulation  and  may  be  synchronized  or  locked  in  by  the  i-f  signal  and  then 
follow  its  deviation.  An  oscillator  remains  in  synchronism  over  a  wider  band,  and  its 
output  is  slightly  greater,  for  high  signal  inputs  than  for  low.  It  has  been  found  to  be 
most  satisfactory  when  operated  at  a  submultiple  of  the  intermediate  frequency.  It  then 
has  the  advantage  of  having  an  output  frequency  which  will  not  feed  back  to  the  previous 
i-f  stages.  However,  it  requires  an  input  signal  above  1  volt  and  in  this  respect  is  no  more 
satisfactory  than  a  grid  limiter.  Discriminators  for  use  with  synchronized  oscillators  must 
be  specially  proportioned  to  take  care  of  the  interaction  with  and  the  loading  of  the  oscil- 
lator circuits. 

A  frequency  detector  which  operates  as  a  locked  oscillator  in  which  the  frequency  is 
controlled  over  the  deviation  range  by  quadrature  feedback  from  the  plate  circuit  of  a 
heptode  tube,  and  which  simultaneously  provides  an  audio  output  in  the  plate  circuit,  has 
been  described.  The  output  is  linearly  proportional  to  the  frequency  deviation  and  is 
independent  of  i-f  amplitude,  provided  the  latter  is  great  enough  to  maintain  synchronism, 

BIBLIOGRAPHY 

Argnimbau,  Discriminator  Linearity,  Electronics,  Vol.  18,  142-146  (March  1945). 

Armstrong,  B.  H.,  A  Method  of  Reducing  Disturbances  in  Radio  Signaling  by  a  System  of  Frequency 

Modulation,  Proc.  I.R.E.,  Vol.  24,  689-740  (May  1936). 
Beers,  G.  L.»  A  Frequency-dividing  Locked-in  Oscillator  Frequency-modulation  Receiver,  Proc.  I.R.E., 

Vol.  32,  730-737  (December  1944). 

Bradley,  W.?E.,  Single-stage  F-m  Detector,  Electronics,  88-91  (October  1946). 
Carnahan,  C.  S.,  and  Kalmus,  H.,  Synchronized  Oscillators  as  F-m  Receiver  Limiters,  Electronics, 

Vol.  17,  108-111,  332-342  (August  1944). 
Crosby,  M.  G.,  Frequency  Modulation  Noise  Characteristics,  Proc.  LR.E.,  Vol.  25,  472-514  (April 

1937). 
Meyers,  S.  T.,  Non-linearity  in  Frequency-modulation  Radio  Systems  Due  to  Multipath  Propagation, 

Proc.  LR.E.t  Vol.  34,  256-265  (May  1946). 
Roder,  H.,  Amplitude,  Phase  and  Frequency  Modulation,  Proc.  I.R.E.,  Vol.  19,  2145-2176  (December 

1931). 

Smith,  D.  B.,  and  Bradley,  W.  E.,  The  Theory  of  Impulse  Noise  in  Ideal  Frequency-modulation  Re- 
ceivers, Proc.  LR.E.,  Vol.  34,  743-751  (October  1946). 
Wheeler,  H.  A.t  Common-channel  Interference  between  Two  Frequency-modulated  Signals,  Proc, 

LR.E.,  Vol.  30,  34-50  (January  1942). 


8-26 


FREQUENCY  MODULATION 


DISTORTION  AND  INTERFERENCE  IN  F-M   SYSTEMS 

By  B.  D.  Loughlin 

In  f-m  systems,  just  as  in  a-m  systems,  distortion  can  result  from  a  non-linearity  of  the 
input-output  characteristic  of  the  modulator  or  frequency  detector.  However,  the  modu- 
lator of  the  transmitter  can  generally  be  properly  designed  to  have  a  substantially  linear 
input-output  characteristic  over  the  operating  range.  In  the  commonly  used  phase 
modulators  which  are  inherently  non-linear  for  large  phase  deviations,  satisfactory  lin- 
earity is  obtained  by  restricting  the  operating  range  to  use  small  phase  deviations  at  the 
modulator,  followed  by  frequency  multiplication  to  obtain  the  desired  frequency  modula- 
tion. It  is  also  relatively  straightforward  to  obtain  linear  frequency  detection  from  the 
commonly  used  f-m  detectors  when  they  are  receiving  an  ideal  f-m  signal  of  constant 
amplitude.  Thus  the  commonly  used  frequency  modulators  and  detectors  are  generally 
designed  so  that  they  contribute  only  a  small  amount  of  distortion  to  the  f-m  system. 

This  section  treats  the  special  lorms  of  distortion  which  are  unique  to  an  f-m  system 
and  which  arise  from  translating  the  f-m  signal  through  the  amplifiers  of  the  transmitter 
and  receiver  and  through  the  transmission  medium.  When  the  f-m  signal  passes  through 
the  amplifiers  of  the  transmitter  or  receiver,  f-m  distortion  can  result  because  of  inade- 
quate band  width,  or  non-linear  phase  characteristic.  Also,  if  the  transmission  charac- 
teristics of  the  amplifiers  are  not  flat  over  the  frequency  deviation  of  the  applied  signal, 
spurious  amplitude  modulation  synchronous  with  the  frequency  modulation  is  introduced, 
which  may  cause  distortion  if  there  is  incomplete  rejection  of  amplitude  modulation  by 
the  f-m  detector  system.  Another  serious  form  of  f-m  distortion,  producing  both  spurious 
amplitude  and  frequency  modulation,  results  from  multipath  transmission  between  the 
transmitter  and  receiver. 


Netwoi* 

characteristics 


a  F-M  DISTORTION  FROM  NON-UNIFORM  AMPLITUDE 
AND  PHASE  CBARACTERISTICS 

When  an  f-m  signal  is  translated  through  an  amplifier  or  network  having  a  non-uniform 
amplitude  or  phase  characteristic,  some  spurious  frequency  modulation  (in  other  words, 

f-m  distortion)  results.  The  f-m  distortion  results 
because  the  various  sideband  components  of  the  f-m 
signal  are  translated  with  different  amplitude  and 
delay  and  thus  do  not  correctly  combine  in  the  out- 
put. The  first  approximation  to  the  distortion  can 
be  obtained  by  a  quasi-steady-state  analysis.  In  a 
quasi-steady-state  analysis,  it  is  assumed  that  at  any 
instant  the  f-m  signal  is  translated  with  an  amplifi- 
cation and  delay  determined  by  the  steady-state 
amplitude  and  delay  characteristics  measured  at  a 
frequency  equal  to  the  applied  instantaneous  fre- 
quency. The  delay  used  is,  of  course,  the  envelope 
delay,  that  is,  the  slope  of  the  phase  characteristic 
at  the  particular  frequency. 

To  illustrate  the  quasi-steady-state  solution,  as- 
sumed amplitude  and  phase  characteristics,  with 
the  resulting  delay  characteristic,  are  shown  in  Fig. 
1.  An  applied  f-rn  signal  with  a  sine  wave  of  modu- 
lation is  shown  together  with  the  resulting  output 
signal  instantaneous  frequency  and  amplitude 
modulation.  It  can  be  seen  that  the  non-uniform 
amplitude  characteristic  introduces  spurious  am- 
plitude modulation  and  no  f-m  distortion  but  that 
the  non-uniform  delay  results  in  an  f^m  distortion 


FIG. 


Quasi-steady-state  Approxima- 
tion for  F-M  Distortion 


<J«e  to  different  delay  for  different  parts  of  the  audio  cycle.  The  resulting  distortion  can 
be  found  by  a  graphical  Fourier  series  analysis,  or  a  Fourier  series  expansion  of  the  output 
f-m  equation.  In  accordance  with  this  approximate  analysis  method  the  output  frequency 
saodnlatkm  would  be: 

f  .m.  =  a  sin  p(t  —  t&)  (1) 

wfeere  a  =  maximum  frequency  deviation,  p  =  angular  modulation  frequency,  and  td  = 
delay  of  circuit  (a  function  of  instantaneous  frequency).  In  this  td  =  io  +  F(a&m  pi), 


8-28 


FKEQUENCY  MODULATION 


Practical  matters,  such  as  ease  of  alignment,  tolerance  of  manufacture,  and  adequate 
selectivity,  frequently  dictate  the  use  of  undercoupled  circuits  in  commercial  f-m  receivers, 
thus  resulting  in  a  non-uniform  amplitude  characteristic.  Even  receivers  designed  to 
have  a  flat  selectivity  curve  are  frequently  in  trouble  as  the  result  of  spurious  amplitude 
modulation  when  the  receiver  is  not  accurately  tuned  or  when  the  set  drifts  out  of  align- 
ment. Thus  it  is  desirable  that  the  f-m  detector  system  have  fairly  complete  rejection  of 
amplitude  modulation  in  order  to  reduce  distortion. 

The  amount  and  type  of  harmonic  distortion  produced  by  the  spurious  amplitude  modu- 
lation is  determined  "by  the  manner  of  the  detector  response  to  it.  As  indicated  by  Figs. 
9(6)  and  9(rf)  (p.  8-23),  the  response  of  the  detector  to  amplitude  modulation  may  be  in 
a  balanced  or  an  unbalanced  manner,  or  a  combination  of  the  two.  As  shown  here,  by 
Fig.  3,  a  receiver  having  a  rounded-top  selectivity  curve,  and  correctly  tuned,  gives 
predominantly  third-harmonic  distortion  when  the  detector  has  a  balanced  response  to 


Detector 
characteristic 
with  balanced 
response  to  a.m. 


j^7  Detector 
y^S       characteristic 
//         with      ' 

^ 


unbalanced 
;poase  to  a_m. 


Frequency  (A/) 


+1.0 


FIG.  3.     Distortion  from  Ampli- 
tude Modulation 


FIG.  4.     Amplitude  Characteristic  for  Two  Cascade 
Double-tuned  Transformers 


amplitude  modulation  and  gives  predominantly  second-harmonic  distortion  when  the 
detector  has  an  unbalanced  response  to  it. 

The  harmonic  distortion  can  be  calculated  by  writing  the  equation  for  the  f-m  detector 
input-output  characteristic  in  terms  of  both  input  signal  frequency  and  amplitude.  Then, 
the  equation  for  the  signal  amplitude  vs.  the  signal  frequency  is  found  from  the  selectivity 
curve.  By  considering  the  input  signal  to  be  frequency-modulated  by  a  sine  wave,  the 
resulting  output  from  the  detector  can  be  written  as  a  trigonometric  series.  Substitution 
of  suitable  trigonometric  expansions  will  give  the  fundamental  and  harmonic  output  signals. 

As  an  example,  consider  a  receiver  with  an  i-f  amplifier  system  including  two  coupled 
circuit  transformers  having  0.7  of  critical  coupling  and  a  band  width  such  that  the  re- 
sponse is  down  6db  at  full  system  deviation  (at  ±75  kc  for  broadcast  frequency  modula- 
tion). Figure  4  shows  that  the  amplitude  characteristic  of  such  an  i-f  system  can  be  closely 
approximated  by  a  parabola,  giving:  Ai  —  1  —  1/2  (A/)2,  where  A/  =  1  corresponds  to  full 
system  deviation.  Now  the  response  of  a  balanced  discriminator  can  be  represented  as 
EG  =  A%k(Af),  where  A%  is  the  applied  signal  amplitude,  and  k  relates  to  the  f-m  detector 
slope.  If  the  amplitude  modulation  of  the  signal  applied  to  the  detector  is  effectively 
reduced  by  some  a-m  reduction  factor  (a),  then  the  amplitude  modulation  due  to  the  i-f 

selectivity  is  effectively  reduced  to  A$  —  1  —  -  (A/)2.    Then  the  detector  output  is: 


£0= 


~  \  (Af)- 


fc  (A/) 


fc  (A/)  -  £  J 


(3) 


Applying  a  sine  wave  of  frequency  modulation,  A/  =  sin  pt  (for  lOOper  cent  modulation), 
then.E©  =  k  sin  pt  -  ^&  sin3  pi  =  k  ( 1  -  '-a)  sinpZ  -f  — -  sin  3pt,  thus  giving:  Per  cent 

£  \  o     /  O 

third-harmonic  distortion  *  — — — -.    For  an  a-m  reduction  factor  (a)  of  0.35,  this 

oU  —  */sa) 
can  be  seen  to  give  approximately  5  per  cent  of  third-harmonic  distortion. 


CROSS-TALK  AND   BEATNOTE   INTERFERENCE  8-29 

It  can  be  seen  from  the  above  that  the  exact  amount  and  the  harmonic  order  of  the  dis- 
tortion will  be  affected  not  only  by  the  selectivity  and  the  manner  and  degree  of  the  f-m 
detector  response  to  amplitude  modulation  but  also  by  the  alignment  of  the  detector 
relative  to  the  intermediate  frequency  and  by  the  tuning  of  the  center  frequency  of  the 
f-m  signal  relative  to  the  center  frequency  of  the  intermediate  frequency.  In  particular, 
it  is  possible  to  obtain  regions  of  high  audio  output  and  large  distortion  when,  as  the  signal 
is  detuned,  the  carrier  level  at  the  detector  system  falls  below  that  necessary  for  good  a-m 
rejection  and  when  the  carrier,  at  the  same  time,  is  tuned  on  the  steep  side-slopes  of  the 
i-f  response.  Such  distorted  side  responses  can  be  considerably  reduced  by  using  a  rounded- 
top  i-f  selectivity  in  conjunction  with  a  well-designed  ratio  f-m  detector. 

10.  DISTORTION  DUE  TO  MULTIPATH  RECEPTION 

When  the  same  radio  signal  is  received  over  several  paths  having  different  delay  times, 
the  several  signals  may  combine  to  give  increased  or  decreased  amplitude  and/ or  an  ad- 
vance or  delay  in  the  resulting  carrier  phase.  For  fixed  differences  in  time  delay  of  the 
paths,  the  relative  phase  of  the  signals  will  vary  with  frequency  of  the  signal.  Thus  multi- 
path  reception  of  an  f-m  signal  will  result  in  spurious  amplitude  and  spurious  phase  modu- 
lation relative  to  the  desired  signal.  The  various  signals  can  combine  so  that  a  substantial 
null  in  transmission  exists  at  certain  frequencies.  As  the  carrier  deviates  through  such  a 
null  frequency,  a  sudden  downward  amplitude  modulation  results  in  conjunction  with  a 
rapid  change  in  phase.  The  sudden  change  in  phase  can  result  in  significant  spurious 
frequency  modulation.  It  appears  that,  where  multipath  transmission  is  expected,  it  is 
of  primary  importance  that  the  f-m  detector  system  have  good  a-m  rejection,  particularly 
in  terms  of  rapidity  of  action  and  amount  of  downward  amplitude  modulation  that  can 
be  accommodated.  In  general,  the  resulting  f-m  distortion  cannot  be  eliminated;  however, 
a  large  amount  of  amplitude  modulation  is  generally  produced  before  such  distortion  is 
severe.  Thus,  good  a-m  rejection  helps  considerably  but  cannot  eliminate  multipath  dis- 
tortion effects. 

Although  numerous  examples  of  multipath  transmission  distortion  have  been  cited  on 
the  higher  frequencies  of  50  to  100  Me,*  it  does  not  appear  to  represent  a  serious  threat  to 
the  f-m  broadcast  industry.  However,  multipath  transmission  distortion  virtually  makes 
voice  communication  by  frequency  modulation  impractical  on  the  long-distance  short- 
wave bands  of  5  to  30  Mc.f  The  many  paths  of  transmission  occurring  during  "skip" 
transmission  on  these  frequencies  result  in  the  familiar  selective  fading  frequently  produc- 
ing serious  f-m  distortion,  particularly  as  the  deviation  is  increased. 

11.  CROSS-TALK  AND  BEATNOTE  INTERFERENCE 

Interference  in  f-m  systems  may  arise  from  other  generated  signals,  such  as  communica- 
tion f-m  or  a-m  signals,  either  received  directly  or  through  spurious  receiver  responses,  or 
it  may  arise  from  noise  signals  of  such  form  as  fluctuation  noise  or  impulse  noise.  The 
response  of  an  f-m  system  to  such  interferences  is,  in  general,  quite  different  from  that  of 
an  a-m  system.  For  example,  cross-modulation  on  amplitude  modulation,  where  the 
modulation  of  a  strong  undesired  a-m  signal  produces  amplitude  modulation  of  a  desired 
signal,  has  no  exact  equivalent  in  an  f-m  system.  The  non-linearities  which  produce  cross- 
modulation  in  amplitude  modulation  produce  some  spurious  signals  in  an  f-m  system  which 
can  have  the  frequency  modulation  of  both  signals,  but  no  direct  cross-modulation  of 
one  carrier  modulation  upon  the  other  carrier  is  produced. $ 

If  two  carrier  signals  of  different  frequency  exist  in  a  linear  system,  the  resulting  signal 
(see  Fig.  5)  has  amplitude  and  phase  modulation  at  the  difference  frequency  rate.  When 
the  ratio  of  the  two  signals  is  substantially  different  from  unity,  the  fractional  amplitude 
modulation  and  the  radian  phase  modulation  are  equal  to  the  fractional  signal  ratio,  that 
is,  the  ratio  of  the  weaker  signal  amplitude  to  the  stronger  signal  amplitude.  The  average 
frequency  of  the  resulting  signal,  being  the  average  number  of  cycles  per  second,  is  the 
frequency  of  the  stronger  signal.  Since  the  instantaneous  frequency  modulation  is  de- 
termined by  taking  the  differential  of  the  phase  modulation,  the  resulting  signal  has  a 
frequency  modulation  that  is  not  only  proportional  to  the  signal  ratio  but  also  directly 

*  Frequency-Modulation  Distortion  Caused  by  Multipath  Transmission,  M.  S.  Corrington,  Proc. 
I.R.E.,  Vol.  33,  878  (December  1945). 

t  Observations  of  Frequency-modulation  Propagation  on  26  Megacycles,  M.  G.  Crosby,  Proc.  I.R.E., 
VoL  29,  398  (July  1941). 

J  Two  Signal  Cross-modulation  in  a  Frequency-modulation  Receiver,  H.  A.  Wheeler,  Proc.  I.R,E., 
Vol.  28,  537  (December  1940). 


8-30 


FREQUENCY  MODtTLATION 


composite  signal 


2  represent  extremes 
mplitude  variations 

1  a  rid  83  represent  extremes 
of  phase  variations 


When  N  small  compared  to  C: 
Peal-  a.m.  (as  a  fraction)  ^i»-j 
Peak  phase  modulation  (radiar 


i)w-g-~& 


proportional  to  the  frequency  difference  between  the  carriers.  Thus,  if  the  two  carriers 
are  applied  to  an  ideal  f-m  detector  system,  the  average  detector  output  will  be  determined 
by  the  average  frequency  of  the  stronger  signal  and  an  f-m  beatnote  will  occur  in  the  de- 
tector output  ha\*ing  a  frequency  equal  to,  and  an  amplitude  proportional  to,  the  differ- 
ence in  frequency  between  the  carriers. 

When  the  ratio  of  the  two  beating  signals  is  far  from  unity,  the  a-m,  p-m,  and  f-m  beat- 
notes  are  all  substantially  sinusoidal.  However, 
as  the  ratio  approaches  unity,  the  beatnote  wave 
forms  depart  from  sinusoidal,  as  shown  in  Fig.  6. 
In  particular,  the  f-m  beatnote  approaches  a  pulse 
wave  form  as  the  signal  ratio  approaches  unity. 

In  an  ideal  f-m  detector  system,  the  detector  re- 
sponds only  to  the  frequency  modulation  of  the 
stronger  signal,  resulting  in  substantially  no  cross- 
talk from  a  weaker  signal.*  This  applies  to  co- 
channel  as  well  as  adjacent  channel  signals,  if  the 
desired-to-undesired-signal  ratio  is  measured  just 
preceding  the  circuit  which  effectively  reduces  or 
which  is  effectively  non-responsive  to  the  amplitude 
modulation  of  the  signal.  For  co-channel  interfer- 
ence, an  f-m  beatnote,  of  variable  frequency,  re- 
sults which  has  considerably  less  amplitude  within 
the  audio  spectrum  (owing  to  the  triangular  f-m 
beatnote  spectrum  as  shown  in  Fig.  5)  than 
would  exist  in  an  a-m  system.  The  direct 
audible  beatnote  interference  does  not  exist  for 
adjacent-channel  interference  in  a  conventional 
f-m  receiver. 

In  practical  f-m  receivers  the  ideal  perform- 
ance in  regard  to  co-channel  and  adjacent- 
channel  interference  is  not  realized  because  of  inadequate  a-m  rejection.  The  amount  by 
which  the  signal  ratio  can  approach  unity  is  generally  limited  by  the  downward  a-m  rejec- 
tion capability  of  the  system.  This  generally  means  that  the  signal  ratio  can  get  to  within 
only  3  to  6  db  of  equality  before  cross-talk  results,  even  with  good  f-m  detectors.  When 
considering  the  signal-to-interference  ratio  for  the  adjacent  channel  case,  the  most  adverse 
condition  during  modulation  must  be  taken.  This  exists  when  the  undesired  signal  has 
maximum  deviation  toward  the  center  of  the  i-f  pass  band,  and,  owing  to  the  sharp  skirt 
selectivity,  this  condition  may  differ  considerably  from  the  unmodulated  case. 

Another  practical  limitation  results  from    Amplitude          s~\  s~*\  /-Xv 

inadequate  rejection  of  superaudibie  ampli- 
tude modulation  produced  in  adjacent- 
channel  interference,  where  the  beatnote 
generally  exceeds  200  kc.  The  grid-bias 
limiter  plus  balanced  discriminator  type  of 

f-m  detector  system  suffers  from  this  limi-    a**^1^0"     \S       ^T 
tation.    The  usual  limiter  grid-circuit  and 
diode  load  time-constants  do  not  permit 
following   of  the   superaudibie   amplitude 

modulation,  resulting  in  an  effective  unbal-    modulation  \^/     \^/  ~\~7 \~f" 

ancing  of  the  discriminator  and  a  change  in  V        V 

average  output  of  the  limiter.  This  limita- 
tion frequently  requires  a  signal-to-inter- 
ference ratio  of  about  20  db  to  eliminate 


/o 
Frequency  of  N  (£) 

FIG.  5.    Addition  of  Two  Carrier  Signals 


Phase     *\ 

roodjuktton      \ 


Frequency 

modulation 


s\ 


Vertical  scale  is  actual  modulation  x  -j- 
FIG.  6.    Beatnote  Wave  Forms 


cross-talk  from  adjacent    channel   signals, 
when  this  f-m  detector  system  is  used.    De- 
tector systems  not  including  the  equivalent  of  this  time-constant  limitation  are  generally  able 
to  tolerate  a  3-  to  6-db  signal-to-interference  ratio  to  eliminate  adjacent  channel  cross-talk. 


12.  FLUCTUATION  NOISE  INTERFERENCE 

Fluctuation  noise,  such  as  thermal  noise  of  resistive  impedances,  and  shot  noise  and 

imaiopitl  noise  of  vacuum  tubes,  can  be  considered  equivalent  to  a  uniform  spectrum  of 

* Coma«saClmiifteJ  Interference  between  Two  Frequency-modulated  Signals,  H.  A.  Wheeler,  Proc. 


IMPULSE  NOISE  INTERFERENCE 


8-31 


energy  in  which  the  components  have  random  phase  or  timing.  When  the  noise  is  small 
compared  to  the  signal,  any  individual  noise  component  will  beat  with  the  carrier  to  give 
an  f-ni  beatnote  as  illustrated  in  Fig.  5.  Thus  the  resulting  audio  noise  consists  of  pre- 
dominantly high  audio-frequency  components  giving  a  characteristic  high-frequency  hiss 
for  f-m  noise  as  compared  to  the  uniform  spectrum  with  considerable  low-frequency 
rumble  for  a-m  noise.* 

To  find  the  rms  value  of  the  audio  noise,  the  output  noise  spectrum  can  be  squared,  the 
resulting  squared  spectrum  can  be  integrated  over  the  audio  band,  and  the  square  root 
of  the  integral  taken.  Using  this  for  the  simple  case  of  an  audio  system  with  uniform 
response  and  sharp  cut-oft,  the  f-m  signal-to-noise  ratio  to  the  a-m  signal-to-noise  ratio 
(called  the  f-m  improvement  ratio)  is  found  to-  be  V§  fd/fa,  when  fa  is  the  maximum  audio 
frequency  and  id  is  the  maximum  frequency  deviation  of  the  system.  For  15-kc  audio 
and  75-kc  deviation  this  gives  an  f-m  improvement  of  18.8  db. 

By  including  a  de-emphasis  low-pass  filter  in  the  receiver,  which  is  compensated  for  by 
a  complementary  pre-emphasis  circuit  at  the  transmitter,  the  f-m  signal-to-noise  ratio 
can  be  further  improved.  In  this  case  the  f-m  signal-to-noise  ratio  including  de-emphasis 
to  the  a-m  signal-to-noise  ratio  not  including  de-emphasis  is  given  by: 

fd 


where  /o  is  the  frequency  for  3-db  attenuation  of  the  de-emphasis  filter.  For  broadcast 
frequency  modulation  with  15-kc  audior  75-kc  deviation,  and  75-microseeond  de-emphasis 
time  constant  (/o  =*  2.12  kc),  this  gives  an  f-m  improvement  of  32  db. 

The  above  relations  are  derived  on  the  assumption  that  the  noise  is  sman  compared  to 
the  carrier  signal.  In  the  region  where  the  peak  noise  is  almost  equal  to  the  peak  carrier, 
the  simple  relations  are  inadequate,  and  actually  the  f-m  improvement  is  rapidly  lost  as 
the  signal  is  made  weaker.  The  approximate  threshold  for  f-m  improvement  is  when  the 
peak  carrier  equals  the  peak  noise. 

Signal-to-noise  ratios  are  frequently  expressed  in  terms  of  the  rms  audio  signal  output 
for  30  per  cent  frequency  modulation  to  the  rms  audio  noise  when  the  carrier  is  unmodu- 
lated. Using  this  definition  and  applying  the  approximate  equa- 
tions at  the  f-m  improvement  threshold  (where  peak  carrier 
equals  peak  noise  after  selectivity),  the  signal-to-noise  ratio  for 
broadcast  frequency  modulation  with  150-kc  i-f  pass  band  is 
about  40  db  at  the  threshold.  Thus,  in  this  case,  the  frequently 
used  30-db  signal-to-noise  ratio  is  near  the  knee  of  the  improve- 
ment threshold,  and  when  the  receiver  has  good  a-m  rejection 
it  is  determined  by  the  signal  level  which  approximately  gives 
peak  carrier  equal  to  peak  noise.  For  broadcast  f-m  receivers 
with  150-kc  i-f  pass  band,  a  300-ohm  antenna,  and  an  assumed 
receiver  noise  factor  of  6  db,  the  threshold  level  (peak  carrier  = 
peak  noise)  is  at  about  104  db  below  1  volt.  This  would  repre- 
sent a  very  well-designed  set.  Normal  design  receivers  have 
an  improvement  threshold  around  90  to  100  db  below  1  volt. 

When  the  peak  carrier  greatly  exceeds  the  peak  noise,  the 
signal-to-fluctuation-noise  ratio  of  an  f-m  system  is  improve*d  by 
using  a  larger  deviation  ratio.  However,  a  small-deviation-ratio 
system  can  have  narrow  receiver  selectivity  resulting  in  less  total  peak  noise  and  thus 
a  lower  threshold  signal  level,  as  illustrated  in  Fig.  7.  Thus,  entertainment  f-m  systems 
where  signal-to-noise  ratio  is  important  are  built  using  a  large  deviation  ratio,  while  com- 
munication networks  where  range  of  coverage  is  important  use  a  small  deviation  ratio. 


Signal  input 


FIG.  7.     Small-  vs.  Large- 
deviation  Ratio  FM 


13.  IMPULSE  NOISE  OTTEBEERENCE 

When  an  impulse,  such  as  automobile  ignition  interference,  is  applied  to  a  receiver,  a 
transient  carrier  pulse  results  having  a  duration  determined  by  the  band  widthf of  the 
receiver  and  a  frequency  determined  by  the  center  frequency  of  the  i-f  selectors.  When 
this  transient  is  added  to  a  desired  carrier,  an  amplitude  and  phase  modulation  of  the 
desired  carrier  results,  depending  upon  the  relative  amplitude,  frequency,  and  phase  of 
the  carrier  and  the  transient.  If  the  carrier  amplitude  is  larger  than  the  peak  amplitude 

*  Frequency-modulation  Noise  Characteristics,  M.  G.  Crosby,  Proc.  I.R.E.,  Vol.  25,  472  (April  1937). 


8-32 


FREQUENCY  MODULATION 


Phase  pufse 
("click") 


Phase  step 
("pop") 


Signal  (Ec)  plus 
impulse  (Ej) 
vectors 


Phase 
modulation 


Frequency 
modulation 


JL 


4- 


of  the  transient,  the  maximum  phase  modulation  that  can  result  is  a  pulse  of  less  than 
1 radian.  In  this  ease  the  audible  interference  produced,  particularly  in  large-devmtion- 

rain'  t'hfcS'ot  mecS  interest,  which  occurs  very  frequently,  the  transient  impulse  ampli- 
tude greatly  exceeds  the  carrier  amplitude.  If  the  desired  carrier  has  a  frequency  equal  to 
the  center  frequency  of  the  selector,  then  the  transient  and  the  carrier  have  a  fixed  phase 
during  anv  one  pulse.  This  results  in  a  pulse  of  phase  modulation  which  can  have  a  max- 
imum phase  displacement  of  approximately  180°  when  the  transient  and  the  desired  car- 
rier  are  almost  out  of  phase.  . 

If  the  desired  carrier  has  a  frequency  different  from  the  center  frequency  of  the  selector, 
then  the  transient  and  the  carrier  will  slip  in  phase  between  the  beginning  and  end  of  any 

one  transient  pulse.  For  certain  con- 
ditions of  starting  phase,  this  case  will 
still  result  in  a  pulse  of  phase  modu- 
lation as  shown  in  Fig.  8.  However, 
there  are  certain  conditions  of  starting 
phase  such  that  the  resulting  signal 
vector  snaps  back  to  its  original  phase 
after  going  through  360°  of  phase  dis- 
placement during  the  pulse  (see  Fig. 
8).  This  produces  a  step  of  phase 
modulation  instead  of  a  pulse  of  phase 
modulation.  Thus,  when  the  desired 
signal  is  not  exactly  on  tune  either  a 
pulse  or  a  step  of  phase  modulation 
can  result  from  a  strong  impulse  noise, 
with  the  probable  occurrence  of  the 
phase  step  becoming  greater  as  the 
signal  is  further  detuned. 

When  a  phase  pulse  is  applied  to 
an  ideal  f-m  detector,  the  output  sig- 
nal is  a  double-polarity  pulse.  This 
double-polarity  pulse  applied  to  the 
de-emphasis  filter  and  audio  system 
results  in  a  unipolarity  pulse  having 
relatively  little  energy  and  a  short 
duration  determined  by  the  cut-off 
frequency  of  the  audio  system.  This 
weak  audio  output  noise  is  sometimes 
called  a  "click."  When  a  phase  step 
is  applied  to  an  ideal  f-m  detector  a  unipolarity  pulse  results.  This  pulse  applied  through 
the  de-emphasis  filter  and  audio  system  gives  a  pulse  with  an  exponential  decay  deter- 
mined by  the  de-emphasis  time  constant  and  thus  having  relatively  more  audio  energy. 
This  louder  audio  noise  is  sometimes  called  a  "pop.**  * 

Thus,  when  a  strong  impulse  is  applied  to  an  f-m  receiver,  either  a  noticeable  pop  or 
weak  click  may  result  in  the  audio  output,  with  the  probable  occurrence  of  the  "pop" 
being  directly  related  to  the  detuning  of  the  desired  carrier  relative  to  the  center  frequency 
of  the  selector.  In  broadcast*  f-m  with  75-kc  deviation  and  75-microsecond  de-emphasis 
time  constant,  the  click  may  have  a  peak  amplitude  between  zero  and  about  6  per  cent 
of  full  modulation,  and  the  "pop"  will  have  a  peak  amplitude  of  about  18  per  cent  of  full 
modulation.  The  amplitude  and  probability  of  occurrence  of  the  pop  is  almost  independ- 
ent of  the  amplitude  of  the  impulse  after  it  exceeds  the  carrier  level  by  several  times. 

To  obtain  the  performance  described  above,  the  f-m  receiver  must  have  good  a-m  rejec- 
tion; otherwise  the  large  amplitude  modulation  resulting  from  the  impulse  noise  will  be 
heard.  Also,  care  must  be  taken  to  see  that  the  receiver  recovers  immediately  after  a 
strong  impulse;  otherwise  the  absence  of  a  signal  immediately  after  an  impulse  may  result 
in  a  large  audio  output  due  to  inadequate  downward  a-m  rejection.  This  trouble  can 
result  particularly  in  a  grid-bias  limiter  with  an  improper  grid  time  constant.  Another 
limitation  preventing  ideal  performance  can  be  spurious  phase  modulation  produced 
within  the  receiver,  during  the  impulse,  from  such  causes  as  change  in  input  capacity  of 
amplifier  tubes. 

*  The  Theory  of  Impulse  Noise  in  Ideal  Frequency-modulation  Receivers,  D.  B.  Smith  and  W  E 
Bwufley,  Pr&c.  I.R.E.,  Vol.  34,  743  (October  1946). 


Audio 
output 


Time — >•  Time — 

FIG,  8.     Impulse  Noise  Interference 


SECTION  9 
PULSE  TECHNIQUES 


PULSES  AND  PULSE  SYSTEMS 
AET  BY  HAROLD  A.  WHEELER  PAGE 

1.  Introduction 02 

2.  Comparison  of  Continuous  Waves  and 

Pulsed  Waves 02 

3.  Types  of  Pulse  Modulation 03 

4.  Speed  of  Information. 03 

5.  Communication 05 

6.  Picture  Transmission 06 

7    Computers 08 

8.  Distance  Measurement 09 

9.  Pulse  Measurements 10 


PULSE  CIRCTnTS 
ART.  BT  J-  J-  OKRENT  PAGB 

10.  Frequency    Multipliers,    Dividers,    and 

Counters 13 

11.  Pulse  Amplifiers , 14 

12.  Pulse  Shaping  Circuits 15 

13.  Relaxation  Circuits 17 

14.  Pulse  Timing  Circuits 19 

15.  Pulse  Modulation  of  an  Oscillator 21 

16.  Modulating  the  Characteristics  of  Pulses  23 

17.  Pulse  Detectors 24 

18.  Vacuum  Tubes 26 

19.  Pulse  Transformers 27 

BY  HAROLD  A.  WHEELER 

20.  Delay  Lines 28 


9-01 


PULSE  TECHNIQUES 
PULSES  AND  PULSE  SYSTEMS 

By  Harold  A,  Wheeler 

1.  INTRODUCTION 

The  various  uses  of  pulses  in  signaling  and  measurements  were  greatly  advanced  during 
World  War  II  with  the  advent  of  numerous  devices  for  aiding  navigation,  -detecting  and 
locating  enemy  craft,  and  performing  difficult  measurements  and  computations.  It  is 
the  purpose  of  this  section  to  give  a  broad  perspective  on  the  many  applications  of  elec- 
tronic pulse  techniques  and  their  limitations,  together  with  a  few  of  the  more  common 
circuits  and  a  large  bibliography  for  further  reference. 

Pulse  coding  is  exemplified  by  the  time-honored  telegraph  codes,  which  were  originally 
operated  slowly  enough  for  crude  mechanical  devices,  manual  transmission,  and  auditory 
reception.  Electronic  pulse  techniques  were  adapted  to  -code  systems  for  amplifying  weak 
signals  and  expediting  the  various  processes. 

Pulsed  radio  waves  date  back  to  the  original  spark  transmitters  of  Hertz  and  others, 
which  set  the  pattern  of  early  radio  communication.  With  the  obsolescence  of  spark 
transmitters  began  the  evolution  of  electronic  pulse  transmitters,  which  had  their  greatest 
use  in  "radar"  during  the  war.  They  now  develop  as  much  as  a  megawatt  of  pulse  power 
at  frequencies  around  3000  megacycles  (wavelength  10  cm) .  In  some  cases,  the  old  rotary 
spark  gap  has  been  revived  to  key  the  new  magnetron  pulse  transmitters. 

2.  COMPAMSON  OF  CONTINUOUS  WAVES  AND  PULSED   WAVES 

Various  kinds  of  information,  such  as  voice  or  music,  are  transmitted  by  corresponding 
modulation  of  a  carrier  wave.  (See  Section  5,  Transients  in  Networks,  and  Section  17, 
Telephone  Systems.)  In  the  simplest  form,  the  carrier  is  a  continuous  wave  of  a  fixed 
frequency,  and  its  amplitude  is  modulated  in  accordance  with  the  sound  wave  or  other 
information  to  be  transmitted.  Amplitude  modulation  is  unique  in  that  the  modulated 
wave  can  be  transmitted  within  the  narrowest  bandwidth  in  the  frequency  spectrum. 
Other  forms  such  as  frequency  modulation  and  pulse  modulation  require  excess  bandwidth 
but  in  return  they  secure  some  advantages  which  may  justify  the  cost  in  bandwidth. 

The  continuous  waves  used  in  amplitude  or  frequency  modulation  on  one  hand,  and 
pulsed  waves  on  the  other  hand,  have  entirely  different  properties  which  require  different 
points  of  view  in  their  application.  These  differences  are  most  pronounced  in  the  selection 
of  one  signal  out  of  several  signals  or  noise  of  comparable  amplitude. 

Figure  1  shows  the  principles  of  selection  between  two  signals,  regardless  of  their  relative 
amplitude.  As  modulated  continuous-wave  signals  are  coextensive  in  time,  frequency 

selection  must  be  used  for  each  channel 

-First channel  by  means  of  band-pass  filters.     Pulsed 

*  j — Second  channel  signals,  however,  can  be  separated  in 

r~ Harmonic  of  second  channel    time,  and  so  it  is  possible  to  use  time 
S~    ~"\  selection  as  well  as  frequency  selection 

\  for  filtering  one  channel  from  another. 

Frequency  or  time   ^ The.  "skirt  ^k^^y"  ^  frequency  se- 
lection   denotes  the    attenuation    just 

FIG.  1.    Selection  in  Frequency  or  Time  outside  the  desired  frequency  band;  in 

time   selection  it  denotes  the  rate  of 

damping  of  one  pulse  to  clear  the  way  for  the  next  pulse  of  another  channel.  Frequency 
selection  is  subject  to  harmonic  interference,  as  shown  in  dotted  lines  in  Fig.  1;  the  anal- 
ogous interference  in  time  selection  is  caused  by  pulse  echoes  in  the  transmission  paths 
in  enclosed  circuits  or  open  space. 

In  any  system  including  several  signals  with  the  same  form  of  modulation,  some  severe 
requirements  have  to  be  met  in  order  to  avoid  interference  between  signals.  Continuous- 

9-02 


SPEED  OF  INFORMATION 


9-03 


wave  signals  require  that  the  response  be  very  nearly  linear  in  order  to  avoid  harmonic 
interference  or  cross-modulation  of  one  signal  by  another.  This  is  not  a  requirement  for 
pulses  separated  in  time,  because  they  are  not  coexistent.  Instead,  the  pulse  signals 
require  that  each  pulse  be  damped  out  immediately  after  its  occurrence  and  that  later 
echoes  be  avoided. 

The  rating  of  equipment  for  pulse  modulation  places  the  emphasis  on  peak  values  rather 
than  average  values.  For  example,  small  vacuum  tubes  can  be  made  to  tolerate  high 
peak  values  of  current  and  voltage  if  they  occur  during  only  a  small  fraction  of  the  time. 
Accumulative  effects,  such  as  heating  and  the  decomposition  of  the  glass,  become  less 
important  because  they  depend  on  average  values. 


(&)   Width  (duration) 


n  _  ru 


)  Spacing  (phase,  frequency) 
FIG.  2.    The  Three  Basic  Types  of  Pulse  Modulation 


. 


3.  TYPES  OF  PULSE  MODULATION 

The  many  possible  ways  of  modulating  pulses  involve  three  basic  types  of  modulation 
as  illustrated  in  Fig.  2.    Height  modulation  (a)  corresponds  to  amplitude  modulation  of 
a    continuous    wave.      Width 
modulation    (6)    and    spacing 
modulation    (c)    involve    only 
the   time  dimensions  and  are 
therefore  not  critically  depend- 
ent on  the  pulse  amplitude. 

The  greatest  advantages  of 
pulses  are  realized  in  time  mod- 
ulation (£>)  or  (c)  as  distin- 
guished from  amplitude  modu- 
lation (a)  in  Fig.  2.  The  tele- 
graph codes  are  an  example  of 
width-and-spacing  modulation. 
It  is  permissible  to  use  ampli- 
tude clipping  or  limiting  cir- 
cuits, since  the  amplitude  need 
not  be  preserved.  Also  the  de- 
tectors are  made  responsive  to  timing  and  can  be  made  unresponsive  to  amplitude  fluctua- 
tions such  as  power-supply  ripple. 

Pulse  echo  systems,  such  as  radar,  utilize  the  timing  of  the  echo  to  determine  the  dis- 
tance. Some  pulse  systems  use  directive  antennas  which  receive  alternately  on  two  crossed 
lobes  of  the  directive  pattern.  In  this  case,  the  relative  amplitude  of  echo  pulses  must  be 
preserved  and  the  direction  of  reception  is  observed  at  the  intersection  of  the  two  lobes 
by  equalizing  the  echo-pulse  amplitudes. 

Some  kinds  of  information,  such  as  numbers,  can  be  transmitted  by  grouping  together  a 
number  of  pulses  in  succession.  Each  group  can  be  evaluated  by  a  pulse  counter.  Mul- 
tiple-pulse coding  is  essentially  similar  to  pulse-width  modulation  but  has  some  advantages 
in  handling  and  in  reliability  of  decoding. 

The  modulation  of  pulses  of  uniform  width  is  similar  to  the  modulation  of  a  subcarrier, 
which  in  turn  modulates  a  carrier.  The  pulse  frequency  is  intermediate  between  the 
modulation  frequencies  and  the  carrier  frequency,  as  is  a  subcarrier  frequency.  In  the 
case  of  pulses,  however,  several  sets  of  short  pulses  of  the  same  frequency  can  be  super- 
imposed for  multiplexing  simply  by  displacement  in  time,  whereas  each  continuous-wave 
subcarrier  would  need  a  different  frequency.  The  pulse  pattern  can  be  subjected  to  any 
method  of  modulation  applicable  to  a  carrier  or  subcarrier,  notably  amplitude  modulation 
as  in  Fig.  2  (a)  and  phase  or  frequency  modulation  as  in  Fig.  2(c). 


4.  SPEED  OF  INFORMATION 

A  time  variation  of  a  quantity  (such  as  current  or  voltage)  may  be  regarded  as  comprising 
a  succession  of  contiguous  pulses  of  varying  amplitude  (see  Fig.  4,  p.  5-28) .  The  speed  of 
information  that  can  be  transmitted  through  a  signal  channel  by  such  a  variation  may  be 
conceived  as  the  maximum  frequency  of  such  pulses  whose  presence  or  absence  can  be 
individually  detected.  (A  space  is  regarded  as  an  absent  pulse,  or  one  of  zero  amplitude.) 
Therefore  the  speed  of  information  is  limited  by  the  frequency  bandwidth.  In  the  case 
of  a  low-pass  channel  (or  one-half  of  a  double-sideband  band-pass  channel),  the  nominal 
minimum  bandwidth  is  one-half  the  maximum  pulse  frequency  as  here  conceived,  but 
somewhat  greater  bandwidth  is  needed  for  insurance  of  pulse  damping  and  for  skirt 


9-04 


PULSE  TECHNIQUES 


selectivity  against  adjacent  frequency  channels.  Figure  3  shows  the  nominal  bandwidths 
required  for  a  speed  of  information  equal  to  2/c.  (The  maximum  pulse  frequency  as  here 
used  is  twice  the  maximum  frequency  of  discrete  pulses  separated  by  intervening  spaces  of 

equal  width;  for  those  separated  pulses 
the  nominal  minimum  bandwidth  be- 
comes equal  to  the  pulse  frequency.) 

The  foregoing  relation  is  based  on 
the  assumption  of  adjacent  non-over- 
lapping pulses,  although  some  overlap 
is  permissible  in  practice.  The  particu- 
lar needs  of  the  system  determine  how 


7"! 

Augmented  bandwidth 

Minimum  bandwidth 

I 
1 
I 

!          i 

G.  3.    Frequency  Bandwidths  Required  for  a  Certain 
Speed  of  Information 


(a) 


(6) 


0   fe         Frequency          f0-fe-*  f0  W0 +/e 

much  the  actual  bandwidth  must  exceed 
the  nominal  minimum  bandwidth.    Fig- 
ure 4  shows  the  distortion  of  a  discrete 
square  pulse  which  is  caused  by  reduction  of  system  bandwidth. 

In  Fig.  4  the  square  pulse  (a)  represents  the  voltage  pulse  or  the  current  pulse  produced 
by  a  system  with  a  very  wide  bandwidth.  In  the  remaining  cases  the  nominal  bandwidth 
of  the  system  (/c)  is  successively  reduced  to  show  its  effect  on  the  output  pulse  caused  by  a 
square  input  pulse.  Case  (6)  shows  a  system  bandwidth  (fc)  approximately  four  times 
that  of  the  nominal  pulse  bandwidth.  This  limitation  causes  sloping  sides  but  retains  a 
flat  top  over  part  of  the  pulse  width.  In  (c),  the  system  bandwidth  is  reduced  to  twice 
the  nominal  pulse  bandwidth,  just  leaving  a  peak  at  the  original  amplitude.  A  system 
bandwidth  equal  to  the  nominal  bandwidth  of  the  square  pulse  (d)  leaves  the  pulse 
slightly  reduced  in  amplitude  and  considerably  widened.  Further,  halving  of  the  band- 
width (e)  reduces  the  amplitude  of  the  output  pulse  to  less  than  one-half  of  the  value  in  (a) 
and  increases  the  width  to  more  than  double.  Cases  (c)  or  (d)  may  be  regarded  as  practical 
compromises. 

Like  frequency  modulation  or  subcarrier  modulation,  pulse  modulation  unavoidably 
increases  the  bandwidth  requirements  for  the  same  speed  of  information,  in  the  manner 
of  the  augmented  bandwidths  indicated  by  dotted  lines  in  Fig.  3.  The  greater  bandwidth 
inherently  increases  the  average  power  of  background  noise  caused  by  thermal  agitation 
of  electrons.  It  also  makes 
possible  a  proportional  in- 
crease in  the  peak  power  by 
pulsing,  while  maintaining 
the  same  average  power. 

If  the  signal  amplitude  is 
comparable  with  the  noise 
amplitude,  a  change  to  puls- 
ing with  its  greater  band- 
width is  no  advantage.  If 
the  signal  is  somewhat 
stronger  than  the  noise, 
however,  and  if  the  pulses 
are  modulated  in  time,  it  is 
found  possible  to  secure  an 
advantage  in  signal-to-noise 
ratio  which  is  comparable 
with  that  obtained  in  wide- 
band frequency  modulation 
over  the  same  bandwidth. 
Therefore,  if  the  augmented 
bandwidth  is  available,  pulse 
modulation  is  another  way 
to  take  advantage  of  it. 

It  may  happen  that,  for 
some    reason,   more   band- 
width is  available  than  the  minimum  needed  for  the  desired  speed  of  information.    At 
very  high  frequencies,  the  accidental  frequency  fluctuations  of  the  signal  may  require  an 
augmented  bandwidth  in  tbe  receiver.    £ome  of  this  excess  bandwidth  may  then  be  utilized 
t0  advantage  by  pulse  modulation. 


FIG.  4. 


The  Widening  of  a  Square  Pulse  by  Reduction  of  Frequency 
Bandwidth 


' — Time 


COMMUNICATION  9-05 


5.  COMMUNICATION 

The  various  types  of  pulse  modulation  have  long  been  used  in  low-speed  and  high- 
speed code  transmission  of  word  messages  (see  Sections  17  and  18,  Telephony  and  Teleg- 
raphy) ,  but  the  greatest  advance  in  pulse  techniques  has  been  utilized  more  recently  in  the 
multiplex  transmission  of  several  voice  channels  on  a  single  microwave  beam  as  a  carrier. 
This  system  is  taken  as  an  example  of  the  communication  possibilities  with  .pulse  modu- 
lation. 

The  carrier  is  modulated  in  short  pulses,  and  the  pulse  spacing  is  modulated  by  sound 
waves.  This  is  the  type  of  modulation  shown  in  Fig.  2(c)  above.  The  multiplex  operation 
is  accomplished  by  interspersed  pulses  as  shown  in  Fig.  5.  A  single  group  of  pulses  com- 
prises a  sequence  including  one 

pulse  assigned  to  each  channel.  1234  1 

One  channel  is  reserved  as  syn-  '     '  *"*  ~  "~ 

chronizing  pulses  to  initiate  each 
counting  sequence  in  reception. 
Each   of    the    remaining    chan-    FIG.  5.     Multiplex  Operation  of  Several  Channels  by  Pulse-time 
nels  is   modulated   by   shifting  Modulation 

its  pulses  in  time  in  accordance 

with  the  sound  wave  to  be  transmitted.  The  amount  of  time  modulation  of  each  pulse  is 
limited  so  that  the  modulation  of  one  pulse  will  never  encroach  on  the  time  allotted  to  the 
modulation  of  the  adjacent  pulses. 

In  the  transmitter,  the  pulses  belonging  to  each  channel  are  synchronized  by  the  first 
channel  but  are  otherwise  separately  generated  and  modulated.  Then  all  channels  are 
combined  with  pulses  interspersed,  and  the  composite  pulse  pattern  is  used  to  modulate 
the  carrier  wave. 

In  the  receiver,  the  modulated  carrier  wave  is  amplified  and  detected,  then  each  sequence 
of  pulses  is  distributed  among  the  several  channels  under  guidance  of  the  synchronizing 
pulses  of  the  first  channel.  The  distribution  of  each  sequence  may  be  accomplished  by 
some  form  of  counting  or  tune  selection.  As  long  as  the  successive  pulses  are  separated 
in  time,  there  is  no  interference  between  channels.  The  time  selection  of  multiplex 
channels  offers  some  advantages  over  frequency  selection,  unless  there  are  strong  echoes 
with  enough  delay  to  overlap  succeeding  pulses,  a  condition  that  can  be  avoided  by  highly 
directive  beam  transmission. 

Reliable  reception  is  generally  possible  if  the  desired  pulse  peaks  are  received  somewhat 
stronger  than  the  peaks  of  noise  or  other  interference.  By  amplitude  limiting  and  clipping, 
the  pulse  peaks  are  flattened  and  the  lower  parts  of  the  pulses  (in  the  noise  background) 
are  discarded.  The  result  is  a  succession  of  square  pulses  with  the  same  timing  as  the  edges 
of  the  received  pulses.  These  reconditioned  pulses  are  distributed  to  the  separate  channels 
for  recovery  of  the  modulation. 

Since  the  sides  of  each  pulse  are  sloping,  the  timing  of  the  reconditioned  pulse  is  still 
subject  to  some  disturbance  by  background  noise,  as  illustrated  in  Fig.  6.  The  noise  causes 

some  vertical  displacement  of  all  parts 
of  the  pulse,  while  there  is  no  change 
in  the  level  which  determines  the  recon- 
ditioning   and    subsequent   detection. 
-  Detection  level     Therefore  the  vertical  displacement  is 
translated  to  a  minor  amount  of  time 
.  Time  displacement,  always  less  than  the  pulse 

width.     The  time  modulation  caused 

FIG.  6.    Pulse  Detection  with  Background  Noise         b^  noise  may  be  compared  to  the  avail- 
able time  modulation  by  the  signal,  to 

determine  its  disturbing  effect.  Increasing  the  frequency  bandwidth  proportionately  in- 
creases the  slopes  and  thereby  decreases  the  response  to  background  noise  (as  in  wide-band 
frequency  modulation) .  Increasing  the  available  time  width  for  modulation  (as  by  decreas- 
ing the  number  of  channels)  decreases  the  ratio  of  the  noise  modulation  to  the  signal 
modulation. 

The  ultimate  effect  of  the  noise  on  the  pulse  slopes  in  Fig.  6  depends  on  the-  kind  of 
detection.  The  simple  detectors  of  time  modulation  operate  on  one  edge  of  every  pulse, 
either  the  leading  or  the  trailing  edge.  Such  detection  retains  the  full  effect  of  the  back- 
ground noise  on  the  sloping  edge  of  each  pulse.  The  time  detection  may  be  designed  to 
operate  on  the  center  of  the  reconditioned  pulse,  in  which  event  there  is  approximate 
cancellation  of  those  noise  components  that  merely  shift  the  pulse  up  and  down,  as  illus- 


9-06  PULSE  TECHNIQUES 

trated  in  Fig.  6,  leaving  only  the  effect  of  those  components  that  distort  the  shape  of  the 
pulse.  The  choice  of  the  kind  of  detection  depends  not  only  on  the  noise  but  also  on  other 
factors  which  may  be  more  important. 

A  common  form  of  interference  in  pulse  transmission  is  echoes  caused  by  reflection  of 
waves  from  objects  in  space  or  from  irregularities  in  transmission  lines.  In  communica- 
tion between  aircraft,  the  principal  cause  of  echoes  is  ground  reflection.  Figure  7  shows 
how  an  echo  may  distort  the  trailing  edge  of  a  pulse.  A  direct  pulse  is  interfered  with  by  a 

slightly  later  echo  pulse  (shown  in 
dotted  lines)  which  is  considerably 
weaker  (below  the  detection  level). 
The  diagram  shows  only  the  envelope 
Detection  level  of  the  puised  wave.  As  the  relative 


—7 
/ 
-^  ---  " 


>~ 


phase  of  the  carrier  wave  of  the  two 

Y  —  ^3»  Time  pulses  may  have  any  angle,  the  echo 

V"**  may  add  or  subtract  on  the  trailing 

FIG.  7.    Pulse  Detection  with  Echo  edge,  as  shown.    Between  fixed  trans- 

mitter and  receiver,  the  effect  of  the 

echo  is  fairly  steady,  varying  slowly  with  frequency  drift  and  environment;  therefore  it 
contributes  little  or  no  disturbance  in  the  receiver.  If  the  distance  is  variable,  as  be- 
tween aircraft,  the  relative  phase  of  the  two  pulses  varies  at  random,  and  so  an  echo 
causes  noise  if  the  time  detection  operates  on  the  trailing  edge. 

Pulsed  waves  are  most  commonly  obtained  by  pulse  modulation  of  a  carrier-frequency 
oscillator  which  delivers  the  required  power  directly  to  the  antenna.  At  the  beginning 
of  each  pulse,  the  oscillation  has  to  build  up  from  the  noise  level.  Therefore  the  oscillator 
acts  as  a  superregenerative  amplifier  of  the  background  noise.  The  resulting  noise  on  the 
leading  edge  of  the  pulse  is  illustrated  in  Fig.  8.  The  modulator  pulse  is  shown  in  dotted 
lines.  At  the  beginning  of  the  modulator  pulse,  the  oscillation  starts  to  build  up  expo- 
nentially from  the  noise  level  and  soon  reaches  equilibrium  at  the  power  level  of  the  oscil- 
lator. However,  the  fluctuation  of  the  noise  causes  a  variable  delay  in  the  build-up  of 
successive  pulses,  which  appears  as  a  "jitter"  in  the  leading  edge.  If  the  time  detection 
operates  on  the  leading  edge,  the  re- 
sult is  noise  in  the  receiver.  This 
effect  is  absent  on  the  trailing  edge 

because  the  latter  is  determined  by  --  1  --  -?/?  ---  ^\  ----  Detection  level 
exponential  damping  from  the  stable 

level  of  the  oscillator  on  the  peak  of     -  1-^2^  ---  ^  -  Time 

the  pulse.     (The  noise  on  the  leading  FIG.  8.    Output  Envelope  of  Pulsed  Oscillator 

edge    can    be    avoided   if  the    pulse 

modulation  is  applied  to  an  amplifier  following  a  continuous  carrier-frequency  oscillator, 
but  this  method  has  other  disadvantages.) 

The  use  of  a  very  short  pulse,  shifted  in  time  with  modulation,  appears  to  be  the  most 
economical  of  power  while  realizing  the  advantages  of  pulse  modulation.  The  pulse  dura- 
tion should  be  nearly  the  least  that  can  be  transmitted  within  the  available  bandwidth 
in  the  frequency  spectrum.  Then  the  available  average  power  can  be  utilized  to  secure 
greatest  pulse  amplitude  for  overcoming  noise.  Since  the  detection  operates  on  the  edges 
of  the  pulses,  greater  pulse  duration  is  no  advantage  and  greater  amplitude  is  a  propor- 
tionate advantage. 

Detection  on  the  leading  edge,  the  pulse  center,  or  the  trailing  edge,  is  a  choice  that 
depends  on  the  nature  of  the  system.  In  beam  transmission  along  a  fixed  path,  echoes  are 
unlikely,  so  detection  on  the  trailing  edge  is  preferable  to  avoid  the  oscillator  noise  on  the 
leading  edge.  Broadcast  transmission,  especially  between  moving  stations,  is  subject  to 
echo  interference,  which  gives  the  advantage  to  detection  on  the  leading  edge;  the  oscillator 
is  then  designed  to  minimize  the  superregenerative  noise.  If  both  the  echoes  and  the 
superregenerative  noise  are  less  than  the  random  background  noise,  center  detection 
would  give  the  best  performance  and  its  complication  might  be  justified. 

The  potentialities  of  multiplex  pulse  transmission  are  indicated  by  the  studies  which 
sfeow  that  the  entire  broadcast  services  for  a  large  city  could  be  transmitted  from  a  single 
microwave  system  centrally  located  on  the  highest  building,  with  a  service  area  limited 
by  tfase  optical  horizon. 

6.  PICTURE  TRANSMISSION 

I®  picture  transmission  by  scanning  methods  (see  Section  19,  Facsimile,  and  Section  2Q, 
km),  pulses  are  relied  on  not  only  for  reproducing  the  picture  elements  but  also  for 
and  synchronizing  the  scanning  process.  Since  the  common  systems  for  picture 


PICTUEE  TRANSMISSION 


9-07 


transmission  utilize  scanning  methods,  many  examples  of  pulse  circuits  and  their  applica- 
tions are  found  in  both  facsimile  and  television. 

Figures  9  and  10  show  respectively  the  essential  components  of  a  picture  transmitter 
and  receiver  with  scanning  by  deflection  of  an  electron  beam,  as  in  present-day  television. 


Line 

Carrier 

frequency 

frequency 

oscillator 

oscillator 

t   Sy 

Line 
"   sawtooth 

<- 

Line 

frequency 
pulse 

-> 

Line  sync, 
pulse 

Camera 

generator 

generator 

generator 

tube 

J,Sc 

^Sy 

Sy 

Picture 

Retrace 

* 

Picture 

-> 

signal       *• 

block-out 
pulse 

Frequency 
divider 

Mixer 

-> 

Modulator 

—  >    Antenna 

generator 

generator 

r 

P 

P 

Sc 

t 

4   Sy 

Sv' 

-  p 

Sy      P         Sy       P 

Frame 

Frame 

Frame 

-  sawtooth 

4- 

frequency 
pulse 

sync, 
pulse 

I 

Sc 

Sy 

Sy 

Picture 

signal 

amplifier 

FIG.  9.    Block  Diagram  of  Picture  Transmitter 


r 

Line 
sawtooth 
generator 

J* 

Picture 

tube                       Projector 

Antenna 

Carrier 
selector 

Det 

Sync. 

Picture                  Picture 

Sc 

and 
amplifier 

separator  —  [ 

reproducer                image 

Frame 
sawtooth 
generator 

P        Sy 

P        Sy            P 

Sy                 Sy          U 

Sc' 

*    P                       P 

Sc 

Picture 

amplifier 

FIG.  10.    Block  Diagram  of  Picture  Receiver 

In  these  block  diagrams,  the  various  functions  are  coded: 
P  =  picture  channel. 

Sc  —  scanning  functions. 

Sy  —  synchronizing  functions. 

The  picture  signal  is  generated  in  the  transmitter  and  converted  back  to  an  image  in 
the  receiver.  The  scanning  function  is  individual  to  tra.nsTn.it.tftr  or  receiver.  The  auto- 
matic synchronizing  operation  is  initiated  in  the  transmitter  by  the  liae-freojiency  oscillator 
and  used  directly  to  time  the  scanning  in  the  camera  tube;  it  is. maintained  by  transmitting 
timing  pulses  along  with  the  picture  signal,  which  are  selected  in  the  receiver  for  holding 
in  step  the  scanning  in  the  picture  tube. 

Figure  11  shows  an  example  of  the  pulses  involved  in  the  scanning  of  a  single  horizontal 
line  in  a  picture.  This  line  is  located  at  the  dotted  line  in  the  pattern  (a).  The  graph,  (6) 
shows  in  the  period  P  the  picture  signal  for  the  line.  As  the  scanning  line  crosses  the 
circular  line,  a  black  pulse  is  generated;  as  it  crosses  each  edge  of  the  black  disk  a  step  is- 


9-08 


PULSE  TECHNIQUES 


(a) 


generated,  two  steps  making  a  wide  pulse.  The  line  signal  is  preceded  and  followed  by  a 
synchronizing  pulse  Sy  which  times  the  successive  lines.  The  sync  pulse  is  communicated 
at  an  "infra-black"  (blacker  than  black)  level  and  so  it  does  not  appear  in  the  retrace 
lines  back  across  the  picture  between  lines.  The  graph  (c)  shows  the  sawtooth  wave  of 
voltage  or  current  which  is  used  to  deflect  the  cathode  ray  or  electron  beam,  in  the  camera 
tube  or  picture  tube,  from  one  end  of  the  line  to  the  other.  ,  ,  . 

The  transmission  of  the  picture  involves  the  communication  of  many  pulses,  each  having 
an  amplitude  proportional  to  the  brightness  of  a  small  element  of  the  picture.  The  "speed 
of  information"  may  be  expressed  as  the  number  of  "independent"  picture  elements  that 
can  be  transmitted  per  second,  or  the  number  of  pulses  per  second.  It  ranges  from  hun- 
dreds in  facsimile  up  to  millions  in 
television.  Completely  independ- 
ent transmission  of  adjacent  pic- 
ture elements  by  successive  pulses 
is  never  attained  because  the  elec- 
trical circuits  and  other  devices 
cause  distortion  resulting  in  over- 
lapping of  adjacent  pulses.  One 
of  the  principal  problems  is  there- 
fore the  design  of  the  circuits  to 
reproduce  the  short  pulses  with 
clean  edges  by  minimizing  ampli- 
tude and  phase  distortion  over 
the  requisite  frequency  bandwidth 
in  the  circuits.  Both  kinds  of  dis- 
tortion are  usually  more  prevalent 
at  frequencies  near  the  limits  of 
fc\  the  frequency  band  required  for 
reproduction  of  the  pulses  and 
lme  steps.  The  higher  frequencies  play 
the  major  part  in  reproducing 
vertical  lines  or  edges;  the  lower 
frequencies,  in  reproducing  long 
pulses  or  background. 


*  "« 

is 


§ 


Levels 
Sync. — 


FIG.  11.    Example  of  Pulse  Functions  in  Picture  Transmission 


The  basic  timing  of  the  system  depends  on  line-frequency  pulses  generated  in  the  trans- 
mitter (Fig.  9)  in  step  with  a  line-frequency  stable  oscillator.  These  pulses  are  used  directly 
to  synchronize  both  the  horizontal  line  scanning  in  the  camera  tube  and  the  sync  pulses 
in  the  composite  signal.  By  means  of  frequency  dividers  or  pulse  counters,  one  of  these 
pulses  is  selected  at  the  proper  time  to  interrupt  the  vertical  scanning  at  the  end  of  each 
frame.  The  resulting  frame-frequency  pulses  are  used  in  a  similar  manner  to  synchronize 
the  vertical  scanning. 

The  synchronizing  pulses  need  to  be  specified  and  preserved  in  shape  only  to  the  extent 
required  by  precision  of  timing  in  the  scanning  operation.  Since  they  are  selected  from 
the  signal  at  a  certain  infra-black  level  which  may  fluctuate  with  picture  content,  it  proves 
necessary  to  hold  the  edge  of  a  sync  pulse  nearly  as  steep  as  the  edge  of  a  step  in  the  picture 
signal.  As  sync  pulses  need  not  be  modulated  in  amplitude,  they  can  be  subjected  to 
limiting  or  clipping  action  to  remove  accidental  changes  in  amplitude.  A  practical  tele- 
vision system  may  have  rather  complex  sync  pulses  for  facilitating  the  separation  of  line 
and  frame  sync  signals,  and  for  minimizing  their  susceptibility  to  interference  from  the 
picture  signal  or  other  disturbances. 


7.  COMPUTERS 

The  solution  of  certain  problems,  such  as  the  trajectory  of  a  missile,  requires  elaborate 
calculations,  which  are  laborious  even  when  done  on  mechanical  computing  machines. 
Electronic  computing  machines  can  increase  the  computing  speed  by  a  factor  of  1000  or 
more,  so  that  a  compilation  of  tables  which  would  take  years  on  a  mechanical  computer 
would  be  done  in  a  few  hours  or  days  on  an  electronic  computer.  The  addition  of  a  digit 
in  electronic  computing  requires  a  few  microseconds  as  compared  with  a  few  milliseconds 
in  mechanical  computing. 

Aa  outstanding  electronic  computer  is  one  of  the  products  of  World  War  II,  the  "elec- 
tronic numerical  integrator  and  computer"  (eniae). 

An  electronic  analog  of  the  mechanical  counter  or  electromechanical  stepping  relay 
may  be  made  up  of  a  number  of  flip-fiop  relaxation  circuits  connected  to  be  switched  con- 


DISTANCE  MEASUREMENT 


9-09 


secutively  by  successive  pulses.  A  single  flip-flop  circuit  is  analogous  to  a  toggle  switch, 
snapping  from  one  condition  to  the  other  and  remaining  there  until  an  external  force  snaps 
it  back.  In  the  block  diagram  of  a  two-digit  ring  counter,  Fig.  12,  each  input  pulse  snaps 
the  "on"  switch  "off"  (to  the  left),  which  in  turn  snaps  the  next  above  switch  "on"  (to 
the  right),  increasing  the  indicated  number  by  one.  If  the  units  column  indicates  nine, 
and  a  unit  is  added,  the  units  "nine"  switch  is  snapped  off,  the  units  "zero"  switch  is 
snapped  on,  and  the  tens  column  receives  a  pulse,  increasing  its  indicated  number  by  one. 
In  the  eniac,  when  going  from  "nine"  to  "zero,"  a  "carry"  switch  stores  the  pulse  for  the 
next  column.  Simultaneous  addition  in  all  columns  is  thereby  made  possible.  The 
"carry"  switch  is  snapped  off  after  addition  in  the  columns  is  complete. 
An  electronic  computer  may  consist  of  the  following: 

1.  A  number  of  counters  to  add  and  store  numbers  represented  by  pulse  groups. 

2.  A  generator  of  suitably  timed  standard  pulse  signals. 

3.  Devices  for  converting  numbers  supplied  to  the  computer  into  pulse  groups  repre- 
senting the  numbers. 

4.  Devices  for  printing  numbers  stored  in  counters. 

5.  Devices  for  combining  the  quantities  in  the  counters  in  different  ways  to  add,  mul- 
tiply, divide,  etc. 

6.  Devices  to  produce  automatic  repetition  of  required  computations,  as  in  numerical 
integration. 

A  basic  decade  ring  counter  was  described  above.  The  decade  counter  may  be  so 
arranged  that,  when  supplied  with  a  standard 
sequence  of  pulses  from  the  timing  generator, 
a  number  of  pulses  corresponding  to  the  number 
stored  in  the  counter  is  transmitted  to  another 
counter  which  adds  it  to  its  own  stored  number. 

Various  groups  of  pulses  from  the  timing  gen- 
erator may  be  selected  and  combined  by  switch- 
ing to  produce  pulse  groups  corresponding  to 
numbers  to  be  supplied  to  the  computer.  The 
numbers  stored  by  a  counter  may  be  indicated 
by  neon  lamps  energized  by  "on"  flip-flops,  as 
indicated  by  the  black  dots  in  Fig.  12.  Card 
punching  or  printing  devices  may  be  similarly 
actuated  to  record  a  stored  number. 

Rapid  switching  is  accomplished  by  means  of 
triple-grid  tubes,  keying  pulses  being  applied  to 
one  control  grid  and  information  pulses  being 
transferred  via  another  control  grid. 

The  flexibility  of  an  electronic  computer  as 
regards  physical  location  of  components,  num- 
ber of  interconnections  possible,  and  the  large 
tolerances  in  amplitude  possible,  facilitates  the  design  of  computers  to  perform  very 
elaborate  computations  with  lightning  speed. 

Where  absolute  numerical  precision  is  not  required,  the  step  counter  may  be  replaced 
by  simpler  devices  relying  on  continuous  integration.  In  one  type,  each  pulse  delivers  an 
incremental  charge  to  a  capacitor  whose  stored  charge  is  indicated  on  a  meter  scale.  In 
another  type,  the  pulse  rate  is  indicated  by  conducting  the  incremental  charges  through  a 
d-c  meter  so  that  the  current  is  proportional  to  the  number  of  pulses  per  second.  Examples 
of  these  types  are  found  in  Geiger  counters,  long  in  use  for  recording  pulses  of  radioactive 
radiation. 

8.  DISTANCE  MEASUREMENT 

Distance  can  be  measured  by  timing  pulses  transmitted  through  a  medium  in  which 
the  wave  velocity  is  known.  In  the  interest  of  precision,  the  duration  of  each  pulse 
should  be  much  less  than  the  time  required  for  the  wave  to  cover  the  distance  in  question. 
The  three  basic  methods  used  in  operating  systems  are  described  in  Section  22. 

Diverse  Waves.  Observations  of  distance  by  this  principle  are  based  on  the  reception 
of  a  pulse  by  two  kinds  of  waves,  over  the  same  distance,  having  different  velocities  so 
that  the  time  difference  in  reception  is  a  measure  of  the  distance.  Examples  of  light  and 
sound  waves  originating  in  simultaneous  pulses  and  traveling  over  the  same  distance  are 
found  in  lightning  and  thunder  or  in  the  seeing  and  hearing  of  a  distant  steam  whistle. 
In  these  cases,  the  only  appreciable  delay  is  in  the  sound  wave,  and  so  the  computation 
of  the  distance  is  based  on  the  velocity  of  sound  in  air.  An  example  of  different  types  of 


Units 
Two-digit  Decade  Counter 


input 


9-10 


PULSE  TECHNIQUES 


waves  in  the  same  medium  is  given  by  the  seismograph  of  an  earthquake.  The  pulses 
are  transmitted  through  the  earth  by  longitudinal  waves  (pressure  waves,  like  sound) 
and  transverse  waves,  having  different  known  velocities  so  that  the  difference  in  time  of 
reception  permits  computation  of  the  distance.  (See  Section  22,  article  9.) 

Pulses  from  Diverse  Locations.  One  of  the  outstanding  radio  navigational  systems 
(loran,  gee,  and  shoran,  Section  22,  article  10)  also  utilizes  pulses  spontaneously  transmitted 
to  the  observer,  but  from  widely  spaced  points  and  carefully  synchronized  in  time.  The 
time  difference  in  their  reception  by  radio  waves  permits  precise  computation  of  the  differ- 
ential distance  relative  to  each  pair  of  spaced  points,  and  such  observations  on  two  different 
pairs  determine  the  position  of  the  receiver. 

Reflection  or  Return  of  Pulse.  The  more  versatile  systems  employ  pulse  waves  trans- 
mitted from  the  observer  to  a  distant  point  and  back  again  to  his  receiver.  The  elementary 

Table  1.    Wave  Velocity 


Kind  of  Wave 

Medium. 

Velocity 

Electromagnetic  (light,  radio)  
Sound       .  .    .        

Free  space  (and  air) 
Air  (atmospheric  pressure,  20  deg  C) 

300  m/ps 
344  m/sec 

Sound  .    ...    .    

Water  (20  deg  C) 

1464  m/sec 

Seismic  (longitudinal,  sound)  .... 
Seismic  (transverse)  

Earth 
Earth 

4-  14  km/sec 
3-  10  km/sec 

sonar  and  radar  systems  utilize  short  pulses  of  sound  or  radio  waves  transmitted  toward 
an  object  and  reflected  back  to  a  receiver  (Section  22,  article  10) .  Special  radar  systems 
for  beacons  or  identification  rely  on  a  pulse  repeater  at  the  object,  which  receives  and  re- 
transmits the  pulses  with  coding  of  some  kind  (radar  beacons,  Section  22,  article  10;  and 
Lanac,  Navar,  and  Teleran,  Section  22,  articles  6  and  11).  In  any  case,  the  round-trip 

time  at  known  velocity  determines 
Table  2.    Characteristics  of  Radar  tne  distance. 

Table  1  shows  various  values  of 
the  wave  velocity  in  different  me- 
diums. Table  2  shows  the  ap- 
proximate range  of  pulse  charac- 
teristics in  radar  systems  of  the 
reflection  type.  Since  the  error  of 
time  measurement  may  be  reduced  to  a  fraction  of  the  pulse  width,  the  corresponding 
distance  error  may  be  reduced  to  the  order  of  1-0.01  mile.  The  range  of  echo  reception 
varies  from  a  few  miles  to  above  200  miles. 


Carrier  frequency  ,        ... 

30-30,000  Me 

Carrier  wavelength  

1  0-0  .  0  1      meters 

Pulse  width 

1  0-0  25      tis 

Pulse  repetition  frequency  

60-4,000    cps 

Pulse  power  

10-1,000    kw 

9.  PULSE  MEASUREMENTS 

PULSE  AMPLITUDE.     A  simple  pulse  of  voltage  or  current  can  be  displayed  on  an 
oscilloscope  and  its  various  properties  determined  from  the  calibration  of  the  scope.    Such 
a  display  is  shown  in  Fig.  13.    The  small  pulses  or 
"pips"  are  superimposed  marker  pulses  carefully 
timed  to  provide  a  time  scale.    A  representative 
scale  might  have  small  marker  pips  every  micro- 
second, and  every  fifth  one  enlarged.    In  Fig.  13  the     '   f    *    *   ' '      \1    t   t    1   t    r    i    r   |  TJme 
pulse  under  observation  is  1  division  wide  at  the    FIG.   13. 
peak  and  about  2  divisions  at  the  base;  it  might  be 
rated  1.5  divisions  wide  at  half  amplitude. 


Oscilloscopic   Observation   of 
Pulse  Characteristics 


A  pulsed  wave  having  many  cycles  of  the  carrier  in  a  single  pulse  presents  a  special 
problem  if  the  scope  cannot  give  a  calibrated  display  of  the  actual  carrier  cycles.  The  wave 
must  be  rectified  for  display,  and  the  performance  of  the  rectifier  is  difficult  to  predict  or 
measure.  If  the  rectifier  responds  quickly,  the  rectified  pulse  may  be  displayed  as  in  Fig. 
13,  which  is  adequate  for  observing  its  width  or  duration. 

An  indirect  method  is  usually  employed  for  rough  measurement  of  the  amplitude  of  a 
repeating  pulsed  wave,  the  pulse  power,  for  example.  The  average  power  is  measured  by  a 
thermal  device  such  as  a  thermocouple  or  bolometer.  The  pulse  duration  is  observed  as 
in  Fig.  13.  Then  the  ratio  of  peak  to  average  power  is  equal  to  the  ratio  of  the  period  of 
repetition  to  the  pulse  duration.  This  method  is  accurate  if  the  time  occupied  by  the  sides 
of  the  pulse  is  much  less  than  the  duration  of  its  peak.  Otherwise  the  pulse  width  is 
indefinite,  usually  approximated  by  the  width  between  the  points  at  one-half  the 


Tjme 


PULSE  MEASUREMENTS  9-11 

power  or  the  peak  amplitude.    (This  same  method  is  applicable  to  simple  pulses  if  there  is 
no  background  of  direct  current  or  voltage,  but  it  is  not  usually  needed  in  this  case.) 

In  the  case  of  a  pulsed  carrier  wave,  some  care  is  required  in  expressing  the  peak  values 
during  the  pulse.  The  peak  power  is  the  mean  power  of  the  carrier  wave  at  the  peak  of 
the  pulse.  If  there  are  minor  ripples  on  the  flat  top  of  a  pulse,  the  pulse  power  is  stated  at 
the  level  of  the  flat  top.  The  peak  voltage  may  be  stated  as  the  peak  value  of  the  carrier 
voltage  at  the  level  of  the  peak  or  flat  top,  since  that  is  the  value  significant  for  voltage 
breakdown.  The  pulse  current  in  a  vacuum  tube  carrying  a  pulsed  wave  is  usually  stated 
as  the  average  value  of  the  current  during  the  peak  or  flat  top  of  the  pulse,  since  that  is 
the  value  readily  measured  by  means  of  an  oscilloscope. 

The  pulse  amplitude  of  a  pulsed  wave  is  best  measured  by  comparison  with  a  continous 
wave  of  known  amplitude,  since  such  a  wave  presents  no  unusual  problem.  The  "notch" 
method  shown  in  Fig.  14  is  based  on  this  principle. 

The  comparison  wave  is  cut  off  or  notched  for  the  i  / \  r  " 

duration  of  the  unknown  pulse.    Then  the  pulsed  \  /    \  / 

wave  is  superimposed  thereon,  and  the  composite i__]L 

wave  is  rectified  and  displayed  on  the  scope.    The    Fl<}_  ^    Mca8Urement  of  Pulse  ^ 

two  amplitudes  are  equalized,  to  give  tne  appearance    tude  by  Comparison   with  Continuous 

of  Fig.  14;  then  the  pulse  amplitude  is  known.    This  Wave 

method  is  independent  of  the  rectifier  and  scope 

characteristics  and  has  been  found  very  useful.    Measurements  have  been  made  as  low  as 

200  MW  peak  pulse  power  and  2  /zw  average  power. 

PULSE  DURATION.  The  simplest  method  of  observing  pulse  width  or  duration  is 
that  shown  in  Fig.  13. 

Another  method  is  based  on  the  frequency  spectrum  of  a  pulse,  illustrated  in  Fig.  15. 
(The  method  of  observing  the  spectrum  is  to  be  described  below.)  The  frequency  spectrum 
of  a  pulse  has  a  width  inversely  proportional  to  the  width  or  duration  of  the  pulse.  Fur- 
thermore, if  the  pulse  has  steep  sides,  the  spectrum  has  a  sharply  defined  minimum  value 
at  a  frequency  differing  from  the  maximum  by  2/c,  as  shown  in  Fig.  15  for  a  pulse  or  a 
pulsed  wave.  The  frequencies  of  minimum  response  can  easily  be  observed  by  a  sharply 
tuned  receiver.  fe  having  been  determined,  the  pulse  duration  is  1/2  /c.  For  example,  fc 

is  1/2  megacycle  for  a  pulse  width 
of  1  microsecond,  and  the  minimum 
response  is  displaced  1  megacycle 
from  the  maximuni  response. 

II,  Mi. ..illll,  Illllllllllllh  illlh.  If  U is ««"«^* to *"«»« ^ 


- — — A  A  -  peak  and  average  values  of  the 

0  2fe  Frequency  /o-2/c-J  f0  L-/0  +  2/C  power  Or  voltage  or  current  of  a 
FIG.  15.  Frequency  Spectrum  of  Pulse  repeating  pulse,  the  duration  may 

be  computed.  The  ratio  of  the 

pulse  duration  to  the  period  of  repetition  is  equal  to  the  ratio  of  average  to  peak  values, 
commonly  called  the  pulse  "duty  cycle." 

AVERAGE  VALUES.  A  repeating  pulse  has  a  definite  average  value  of  power  or 
voltage  or  current.  Any  one  of  these  average  values  may  be  significant  in  determining  the 
heating  or  other  accumulative  phenomena  in  a  circuit  or  vacuum  tube.  For  voltage  or 
current,  it  is  important  to  specify  whether  the  average  or  root-mean-square  value  is  to 
be  measured. 

The  average  power,  or  the  rms  value  of  voltage  or  current,  can  be  measured  by  a  thermal 
instrument  with  a  time  constant  much  greater  than  the  period  of  repetition  of  the  pulses. 
The  peak  voltage  during  the  pulse  is  abnormally  large  for  the  usual  instrument  of  this 
type,  and  so  voltage  breakdown  may  occur,  in  which  event  this  defect  may  be  corrected 
by  redesign. 

The  thermal  instruments  in  common  use  include  the  thermocouple,  the  bolometer 
bridge,  the  lamp  with  photometer,  and  the  resistor  with  calorimeter.  The  most  severe 
requirements  are  met  in  measuring  pulsed  waves  of  ultra-high  carrier  frequencies,  say  1000 
megacycles  and  upward. 

The  bolometer  bridge  is  a  d-c  four-arm  bridge  with  a  temperature-sensitive  resistor  in 
one  arm  which  receives  the  pulse  power  to  be  measured.  This  device  has  been  improved 
by  the  use  of  a  composition  resistor  very  sensitive  to  temperature,  called  the  "thermistor." 
Since  it  is  desirable  for  the  measuring  circuit  to  present  constant  resistance,  the  bridge  is 
rebalanced  by  decreasing  the  d-c  level  in  all  arms  to  restore  the  variable  resistor  to  the 
same  temperature,  establishing  a  d-c  calibration  of  the  power  sensitivity  in  normal  opera- 
tion. The  device  is  sensitive  to  low  power  of  the  order  of  1  milliwatt  and  also  can  be 
adapted  to  greater  power. 


9-12  PULSE  TECENIQTJES 

The  lamp  is  useful  for  medium  power  of  the  order  of  1  watt.  The  radiation  from  the 
lamp  is  indicated  by  a  nearby  photocell  and  meter.  It  has  two  great  advantages :  it  requires 
no  electrical  connections  between  the  pulse  circuit  and  the  indicating  circuit,  and  it  can 
be  calibrated  by  direct  current.  Its  one  great  disadvantage  is  its  large  variation  of  resist- 
ance at  the  high  temperatures  incidental  to  radiation  of  light. 

The  calorimeter  is  useful  for  high  power  of  the  order  of  1  kilowatt.  It  can  be  designed 
for  nearly  constant  resistance  at  very  high  frequencies,  since  resistance  variation  is  not 
essential  in  its  operation  at  moderate  temperatures. 

A  possible  alternative  to  the  thermal  meter  is  a  square-law^  rectifier,  which  may  be 
approximated  by  proper  design  of  a  vacuum-tube  rectifier.  It  is  important  to  hold  the 
peak  value  within  the  range  of  square-law  operation,  which  severely  limits  the  utility  of 
this  type  of  instrument. 

The  average  value  of  current  or  voltage  of  repeating  pulses,  assuming  zero  between 
pulses,  can  be  measured  in  an  ordinary  magnetic  d-c  meter  having  a  time  constant  much 
greater  than  the  period  of  repetition.  In  a  voltmeter,  care  must  be  taken  that  the  pulse 
voltage  is  not  excessive. 

PULSE  FREQUENCY.  Repeating  pulses  commonly  have  a  steady  value  of  the  pulse 
repetition  frequency  (prf).  However,  a  receiver  or  a  replying  transmitter  may  have  a 
random  pattern  of  repetition,  depending  on  the  traffic,  and  then  it  may  be  desirable  to 
have  a  continuous  indication  of  the  average  pulse  frequency  over  a  short  period  such  as 
1  sec  in  order  to  be  aware  of  overloading. 

A  constant  pulse  frequency  is  easily  observed.  It  is  usually  an  audible  frequency  and 
can  be  compared  with  a  calibrated  audio  frequency.  The  integrating  type  of  frequency 
meter  which  is  used  for  direct-reading  audio-frequency  meters  may  be  designed  for  pulse- 
frequency  measurements.  It  gives  the  continuous  indication  desirable  for  monitoring  a 
varying  pulse  frequency.  If  the  pulse  amplitude  and  width  are  uniform,  the  same  result 
can  be  obtained  by  inserting  a  d-c  or  average-power  meter  in  the  circuit  where  it  will 
indicate  an  average  value  proportional  to  the  pulse  frequency. 

PULSE  DETAILS.  A  critical  analysis  of  the  details  of  a  repeating  pulse  or  group  of 
associated  pulses  may  require  a  display  on  the  oscilloscope  with  a  greatly  expanded  time 
scale.  For  example,  the  entire  width  of  the  scale  may  be  only  a  few  microseconds,  even 
though  the  pulse  groups  may  be  separated  by  a  millisecond. 

The  synchroscope  is  a  special  oscilloscope  designed  for  this  purpose.  Each  trace  or 
sweep  is  triggered  by  the  first  edge  of  the  pulse  pattern  to  be  observed,  and  so  close  registry 
of  successive  traces  is  assured,  hence  the  name  "synchroscope.'*  At  the  end  of  each 
trace,  the  spot  waits  for  the  signal  to  start  the  next  trace.  Care  is  taken  to  insure  that  each 
trace  starts  at  the  same  point  and  proceeds  at  the  same  rate.  Any  failure  of  registry  then 
indicates  variations  in  the  pattern. 

FREQUENCY  SPECTRUM.  A  simple  pulse  or  a  pulsed  wave  has  a  frequency  spec- 
trum as  illustrated  in  Fig.  15,  the  former  centered  on  zero  frequency  and  the  latter  on  the 
carrier  frequency.  The  significance  of  the  spectrum  is  the  ability  to  excite  a  circuit  which 
selects  a  bandwidth  much  less  than  the  width  of  the  spectrum. 

The  spectrum  analyzer  is  a  device  for  displaying  the  frequency  spectrum  on  the  scope 
in  the  form  shown  in  Fig.  15.  As  the  trace  progresses  horizontally  along  the  frequency 
axis,  a  narrow-band  receiver  is  tuned  over  the  frequency  range.  The  repeating  pulse  is 
applied  to  the  receiver,  and  the  rectified  output  is  shown  by  vertical  deflection.  The  pulses 
therefore  appear  as  successive  vertical  lines  whose  heights  show  the  relative  amplitude  of 
the  frequency  spectrum.  The  spectrum  is  a  measure  of  the  frequency  bandwidth  required 
for  reproduction  of  the  pulse,  as  well  as  the  interference  that  may  be  caused  in  adjacent 
frequency  channels.  In  the  case  of  a  pulsed  wave,  a  symmetrical  spectrum  indicates  pure 
amplitude  modulation  free  of  frequency  modulation. 

As  appears  from  the  width  of  the  frequency  spectrum,  a  pulsed  wave  does  not  give  a 
sharp  indication  of  resonance  in  a  sharply  tuned  wavemeter.  The  carrier  frequency  is  best 
defined  by  the  center  of  a  symmetrical  spectrum.  Therefore  it  is  customary  to  provide 
the  analyzer  with  a  very  sharply  resonant  calibrated  circuit  whose  frequency  can  be 
adjusted  to  put  a  narrow  gap  in  the  center  of  the  spectrum  and  thereby  to  determine  the 
carrier  frequency.  The  calibration  of  the  trap  circuit  is  made  on  the  narrow  spectrum  of  a 
continuous  wave  or  long  pulses  of  known  carrier  frequency. 

(For  oseiHoscope  technique,  see  Section  11,  Wave  Analysis,  and  Section  20,  Television.) 


FBEQUENCY   MULTIPLIERS,    DIVIDERS,    AND    COUNTERS      9-13 


PULSE  CIRCUITS 

By  J.  J.  Okrent 

10.  FREQUENCY  MULTIPLIERS,  DIVIDERS,  AND  COUNTERS 

The  abrupt  changes  in  amplitude  associated  with  pulse  phenomena  give  rise  to  a  wide- 
band frequency  spectrum,  as  indicated  above  (see  Computers,  article  7  above;  also  Section 
11,  Frequency  Measurements).  The  frequency  spectra  of  certain  periodic  wave  forms,  as 

rlfuuLrLn_ 


+B 


FIG.  1.    Counting  Type  of  Frequency  Divider 


expressed  by  Fourier  series,  are  well  known.  Periodic  pulses  have  strong  components  of 
high-order  harmonics.  These  are  produced  for  frequency  comparison  by  making  a  stable 
sinusoidal  oscillator  synchronize  a  relaxation  circuit  which  produces  the  high-order  har- 
monics of  the  stable  frequency.  The  harmonics  are  compared  (by  zero  beat,  for  example) 
with  a  signal  whose  frequency  is  to  be  standardized.  Substantial  output  of  a  harmonic 
frequency  is  obtained  by  making  the  relaxation  pulse  excite  a  circuit  sharply  resonant  at  a 
chosen  harmonic  frequency.  Pulse  circuits  incidentally  radiate  interfering  power  at 
harmonic  frequencies  unless  they  are  adequately  shielded  and  filtered. 

Frequency  division  may  be  accomplished  by  synchronizing  a  relaxation  oscillator  at  an 
integral  submultiple  of  the  synchronizing  signal  frequency.  The  natural  period  of  the 
relaxation  oscillator  is  made  somewhat  longer  than  the  interval  occupied  by  the  selected 
number  of  synchronizing  pulses,  and  so  the  pulses  expedite  the  relaxation  in  each  cycle. 

Another  method  of  frequency  division  utilizes  a  "ring-counter"  circuit  which  counts 
the  required  number  of  pulses  and  then  generates  a  trigger  pulse  and  starts  counting  over 
again.  In  Fig.  1,  successive  pulses  of  current  /i  increase  the  charge  on  the  capacitor  £3 
and  thereby  its  voltage  E%,  for  each  pulse  until  the  output  thyratron  conducts,  discharging 
Cz  and  producing  an  output  pulse. 


-fB 


Output 


•f  Bias 


Output 


Plate 
of  B-2 


Plate 
of  B-l 


J 

J 


Plate 
of  B-0 


Input 


Input 


FIG.  2.    Flip-flop  Circuits  in  a  Counter 

Figure  2  shows  flip-flop  circuits  of  a  type  used  in  counters  and  computers.    Successive 
input  pulses,  not  necessarily  at  regular  intervals,  switch  successive  flip-flops  to  an  indicate 


9-14 


PULSE  TECHNIQUES 


ing  position.  In  each  flip-flop,  either  triode,  A  or  B,  may  conduct;  the  other  triode  is  then 
cut  off.  Also,  if  triode  A-0  is  cut  off,  triodes  A-l  and  A-2  are  conducting.  A  positive  input 
pulse  will  then  switch  triode  .4-0  into  conduction,  and  the  coupling  between  the  plate  of 
B-Q  and  the  grid  of  B-l  will  switch  triode  A-l  into  conduction.  The  control  grid  of  triode 
.4-1  is  thus  made  sensitive  to  the  next  input  pulse,  and  the  high  plate  voltage  on  B-l  may 
actuate  for  that  digit  an  indicating  device  such  as  a  neon  lamp.  The  coupling  between 
triodes  #-2  and  B-0  completes  one  ring,  so  that  the  cycle  is  repeated  every  three  input 
pulses.  Output  pulses  with  a  frequency  one-third  that  of  the  input  pulses  are  thereby 
obtained. 

11.  PULSE  AMPLIFIERS 

The  requirements  to  be  met  by  pulse  amplifiers  are  generally  similar  to  those  of  television 
carrier-frequency  and  video-frequency  amplifiers  (see  Section  7,  Wide-band  and  I-f  Am- 
plifiers). The  amplifiers  must  pass  the  essential  frequency  components  of  the  signal  to 
be  amplified,  with  uniform  gain  and  time  delay.  Any  excess  bandwidth  passes  needless 
background  noise.  A  compromise  choice  made  in  radar  systems  allows  a  video  bandwidth 
(in  megacycles  per  second)  equal  to  the  reciprocal  of  the  pulse  duration  (in  microseconds) , 
and  double  this  bandwidth  for  the  modulation  sidebands  of  a  pulsed  carrier.  This  is 
twice  the  nominal  minimum  bandwidth  defined  on  p.  9-11.  Some  additional  carrier- 
frequency  bandwidth  is  added  for  tolerance  of  detuning  from  various  causes  such  as 
frequency  drift. 

The  amplitude-and-phase  spectrum  concept  (see  Section  5,  article  13)  is  useful  in  specify- 
ing low-pass  and  band-pass  amplifiers  for  pulses  and  pulse-modulated  carrier  waves. 

Fast  recovery  of  normal  operation  after  overloading  by  strong  signals  is  required  in 
systems  like  radar,  in  which  a  weak  pulse  may  immediately  follow  a  strong  pulse.  Figure 
3  shows  an  i-f  amplifier  stage  having  quick  recovery.  The  grid  bias  returns  to  normal 


VvA, t-B 


FIG.  3.    Intermediate-frequency  Amplifier 


immediately  after  a  strong  pulse  because  there  is  negligible  d-c  resistance  between  grid 
and  ground,  and  the  cathode-circuit  time  constant  is  only  0.1  microsecond.  The  cathode- 
circuit  time  constant  is  made  the  minimum  consistent  with  sufficient  bias  and  bypassing. 
Plate  current  is  supplied  through  the  damping  resistor  for  the  stage.  Fast  recovery  of 
normal  plate  voltage  after  a  strong  pulse  is  relatively  unimportant  because  the  operation 
of  a  pentode  tube  does  not  depend  critically  on  plate  voltage.  The  inductance  L  is  made 
to  resonate  the  distributed  capacitance  of  the  wiring  and  tubes,  Cd,  without  any  added 
lumped  capacitance,  in  order  to  obtain  maximum  stage  gain  with  the  required  bandwidth. 

Video  pulse  amplifiers  are  frequently  specified  in  terms  of  the  permissible  distortion 
of  rectangular  pulses  by  the  amplifier,  rather  than  by  amplitude  and  phase  characteristics. 
Thus,  it  may  be  required  that  a  specified  input  pulse  after  amplification  shall  have  a  cer- 
tain maximum  rise  time,  fall  time,  ripple  ratio,  etc. 

Resistance-coupled  stages  are  used  for  voltage  amplification.  Because  high  video  fre- 
quencies are  usually  involved,  high-transconductance  pentodes  and  relatively  small 
coupling  resistors  are  used.  The  product  of  the  coupling  resistance  by  the  shunt  capaci- 
tance places  a  lower  limit  on  the  time  of  rise  and  fall  "of  output  pulses.  Shunt  capacitance 
of  un-bypassed  components  and  wiring  should  be  made  as  small  as  practicable. 

In  Fig.  4,  an  input  pulse  having  zero  rise  time  would  produce  at  the  f oUowing  grid  an 
amplified  pulse  having  a  leading  edge  rising  exponentially  with  a  time  constant 

n  +  ^  ("*  ohms,  microfarads,  and  microseconds).    The  amplified  pulse  would  reach  00 

per  ©eat  of  peak  amplitude  in  2.3  times  the  time  constant.  The  plateau  of  the  pulse 
across  TI  decays  exponentially  with  the  time  constant  fo  +  r2)Cc,  ordinarily  much  longer 


PTJLSE  SHAPING  CIRCUITS 


9-15 


than  T,  the  duration  of  the  pulse.  If,  for  example,  less  than  1  per  cent  decay  is  permissible 
from  this  coupling  alone,  make  fa  -f  r£Cc  greater  than  100T. 

The  high-frequency  response  of  resistance-coupled  amplifiers  may  be  improved  by 
compensation  and  filter  techniques  discussed  elsewhere  in  this  book  (see  Section  7,  Wide- 
band Amplifiers,  and  Section  20,  Television) . 

When  power  amplification  or  impedance  change  is  required,  transformer  coupling  may 
be  used.  High-power  pulses  are  economically  obtained  from  tubes  having  low  average- 
power  ratings  by  permitting  space  current  to  flow  only  during  pulses.  The  control  grid 


+  SC  +B  +B 

Amplifier  Cafhode  Follower 

FIG.  4.    Video-frequency  Amplifier 

in  a  high-power  amplifier  is  therefore  biased  negative  and  driven  positive  during  the 
pulses.  Transformers  may  be  used  to  provide  the  polarity  inversion  which  may  be  re- 
quired in  successive  high-power  stages  and  also  to  permit  impedance  matching.  Pulse 
transformers  will  be  described  below. 

Cathode  followers  (see  Fig.  4)  are  useful  in  obtaining  high-voltage  positive  pulses  across 
low  impedance  such  as  the  coaxial  lines  commonly  used.  In  successive  stages  of  high- 
power  pulse  amplifiers,  cathode  followers  make  transformers  unnecessary  for  polarity 
inversion.  However,  transformers  are  necessary  if  voltage  amplification  is  required  in 
addition  to  the  current  amplification  obtained  in  a  cathode-follower  stage. 


12.  PULSE  SHAPING  CIRCUITS 

Many  of  the  wave  forms  used  in  pulse  work  are  obtained  by  means  of  exponentially 
changing  voltages  and  currents  and  by  clipping  or  limiting  action  in  the  tubes.  In  sharp 
contrast  with  linear-amplifier  practice,  the  operating  conditions  of  the  tubes  in  pulse 
shaping  circuits  are  so  chosen  that  grid  current*  may  flow  or  plate  current  may  be  cut  oft 
in  order  to  distort  a  pulse  or  to  reconstruct  a  clean  pulse. 

Clipping  or  Squaring.  In  Fig.  5,  a  sinusoidal  voltage  is  applied  to  the  grid  of  a  tube 
through  a  high  resistance.  As  the  grid  goes  positive,  the  grid-to-cathode  resistance  falls 


FIG.  5.    Squaring  a  Sine  Wave 


abruptly,  flattening  the  positive  peak  of  grid  voltage.  As  the  grid  goes  negative,  the  plate 
current  is  cut  off,  flattening  the  negative  peak.  The  output  wave  form  is  approximately 
square  and  can  be  squared  further  by  added  stages. 

Pulse  Narrowing.  In  Fig.  Qa  the  time  constant  r\Ci  is  short  as  compared  with  the 
duration  but  not  as  compared  with  the  time  of  rise  and  fall  of  the  applied  pulse,  e\.  The 
voltage  across  r\t  from  the  step  at  each  edge  of  the  pulse,  decays  exponentially.  The  pulse 
is  said  to  be  differentiated,  because  the  voltage  across  n  is  approximately  proportional  to 
the  derivative  of  the  applied  pulse  voltage.  Either  the  leading  or  the  trailing  edge  may  be 
used  by  the  following  stage.  If  the  bias  on  the  tube  Tz  is  large  enough  to  cut  off  the  plate 


9-16 


PULSE  TECHNIQUES 


current,  only  the  positive  (leading)  impulse  is  amplified.    If  the  bias  is  zero  and  r2  is  large, 
only  the  negative  (trailing)  impulse  is  amplified. 

Long  pulses  may  be  shortened  to  a  predetermined  duration  by  application  through  a 
high  resistance  (in  Fig.  6b  the  plate  resistance  of  the  pentode)  to  a  delay  line  which  is  in 


fB         Bfas 
FIG.  6a.    Narrowing  a  Pulse  by  R-c  Differentiation 

parallel  with  a  resistance  equal  to  the  characteristic  impedance  ZQ  of  the  line.  The  far 
end  of  the  line  is  short-circuited,  so  that  the  pulse  at  the  input  end  of  the  line  is  canceled 
by  a  reflected  pulse  of  opposite  polarity  in  twice  the  one-way  delay  time,  td,  of  the  line. 
The  undesired  pulse  of  reverse  polarity  which  occurs  at  the  end  of  the  input  pulse  is 
clipped  in  a  succeeding  circuit. 


2** 


utput 


Input 


FIG.  6£.    Narrowing  a  Pulse  by  Use  of  a  Delay  Line 

Another  arrangement  for  obtaining  a  short  pulse  of  predetermined  duration  from  a 
long  pulse  is  shown  in  Fig.  6c.  A  half  sine  wave  of  short  duration  is  produced  in  the  plate 
circuit  by  the  application  of  a  long  pulse.  The  period  of  the  half  sine  wave  is  determined 
by  the  inductance  of  the  coil  and  the  distributed  capacitance  across  it.  The  diode  damps  the 
oscillation  on  the  second  half  cycle,  reducing  the  oscillation  after  the  first  half  cycle  to  a 
negligible  level. 

+  B 


FIG.  6c.    Narrowing  a  Pulse  by  Use  of  an  Oscillatory  Circuit 

Pnlse  Widening,  Integration  is  said  to  occur  in  Kg.  7.  A  positive  pulse  on  the  grid 
of  tube  A  quickly  discharges  capacitor  Ci,  which  recharges  exponentially  with  the  time 
constant  rid.  Resistance  r2  is  assumed  large  as  compared  with  r^  and  rsCa  large  as  com- 
pared with  rid.  Tube  B  is  cut  off  during  enough  of  the  exponential  discharge  to  provide 
tl*«  required  pulse  duration.  The  wave  forms  in  solid  line  are  typical  of  the  circuit  shown. 
IV  eoaanectiAg  rs  to  a  positive  voltage  instead  of  K>  ground,  a  shorter  duration  and  time 
ol  fall  is  obtained,  as  shown  by  the  wave  forms  in  broken  line. 


RELAXATION  CIRCUITS 


9-17 


+  B 


+  B 


A— -^- 

^2  —  connected  to  +  B 
instead  of  to  ground 

FIG.  7.    Widening  a  Pulse 

Clamping  or  D-c  Reinsertion.  When  it  is  required  that  the  baseline  of  a  pulse  wave 
form  remain  at  a  fixed  voltage  in  spite  of  capacitive  coupling  and  changing  wave  form,  the 
arrangement  shown  in  Fig.  8  may  be  used.  If  the  two  wave  forms  shown  are  coupled  by 
capacitor  C  and  resistor  r\  without  the  diode,  the  voltage  limits  vary  as  shown  because 
the  average  voltage  must  remain  zero.  The  use  of  a  diode  results  in  a  short  coupling  time 
constant  for  negative  voltages,  so  that  the  wave  extends  almost  wholly  in  the  positive 
direction.  The  resistance  of  n  is  much  greater  than  the  resistance  of  the  diode  in  the  eon- 
ducting  direction  (ra).  If  no  grid  current  flows,  the  rectifying  action  of  the  diode  main- 
tains on  the  capacitor  a  sufficient  charge  to  hold  substantially  the  entire  wave  positive 


,jL47m_ 


FIG.  8.    Baseline  Clamping 


relative  to  ground.  The  diode  also  permits  fast  recovery  of  normal  bias  if  grid  current  is 
drawn  by  a  large  positive  pulse,  because  a  negative  voltage  is  quickly  discharged  through 
the  diode. 

13.  RELAXATION  CIRCUITS 

Relaxation  circuits  are  oscillators  in  which  little  if  any  energy  is  stored  from  one  relaxa- 
tion cycle  to  the  next.  The  circuits  have  two  conditions  in  which  they  are  at  least  tem- 
porarily stable.  When  one  condition  becomes  unstable,  the  oscillator  shifts  abruptly  to 
the  other  condition. 

An  example  of  a  circuit  which  has  two  permanently  stable  conditions  is  shown  in  Fig.  9. 
One  tube  is  conducting  and  the  other  cut  off  in  each  of  the  stable  conditions.  Conduction 


FIG.  9.    Flip-Sop  Circuit  Stable  in  Either  Condition 

is  switched  back  and  forth  by  successive  trigger  pulses,  hence  the  designation  "flip-flop" 
circuit.    Typical  voltages  are  shown  in  the  figure. 


9-18 


PULSE  TECHNIQUES 


In  Fig.  10,  a  capacitor  replaces  one  of  the  plate-to-grid  coupling  resistors,  and  direct 
coupling  is  used  between  the  other  plate  and  grid.  Plate  current  cut  off  in  tube  T-2  and 
conduction  in  tube  T-3  make  a  permanently  stable  condition.  A  trigger  pulse  through 
tube  T-l  makes  tube  T-2  conduct  and  tube  !T-3  be  cut  off  in  a  temporarily  stable  condition 


T-I 


FIG.  10.    Triggered  Multivibrator 


which  ends  when  C  discharges  sufficiently  through  r.  The  circuit  then  switches  back  to 
the  permanently  stable  condition.  The  circuit  is  known  as  a  triggered  or  one-pulse  multi- 
vibrator, or  a  '"univibrator." 

The  circuit  shown  in  Fig.  11  has  only  temporarily  stable  conditions  and  therefore  runs 
free.    It  may  be  synchronized  at  the  trigger  frequency  or  integral  submultiples  thereof. 


FIG.  11.    Free-running  Multivibrator 

A  transformer  is  used  (instead  of  a  second  tube)  for  feedback  polarity  reversal  in  the 
one-tube  blocking  oscillator  circuit  shown  in  Fig.  12.  It  is  free  running  unless  the  C  bias 
is  increased  to  limit  the  oscillation  to  one  pulse.  A  temporarily  or  permanently  stable 
condition  exists  between  pulses  while  the  grid  capacitor  is  charged  negative  beyond 
cutoff.  The  other  extreme  condition  exists  during  the  pulse  while  the  rate  of  increasing 


Output  >, 


-C 


-C    •="  ~ 

FIG.  12.    Blocking  Oscillator 


plate  current  is  limited  by  the  inductance  in  the  plate  circuit;  this  condition  causes  grid 
current  to  charge  the  grid  capacitor  ready  for  the  blocking  period.  Because  plate  current 
Sows  only  during  the  pulses,  the  blocking  oscillator  works  economically  with  the  low  im- 
pedances and  high  currents  necessary  for  short  pulses  across  the  inherent  capacitance. 

ing  discharge  tubes,  such  as  thyratrons  and  gas  diodes,  are  used  in  generating 
pulses  and  sweep  voltages  by  periodic  discharge  of  a  capacitor  each  time  the 


PULSE  TIMING  CIRCUITS  9-19 

capacitor  is  charged  to  the  ionizing  potential  of  the  discharge  tube.    The  capacitance  and 
the  charging  impedance  are  chosen  to  give  the  desired  period  of  pulse  repetition. 

A  number  of  relaxation  circuits  particularly  adapted  for  different  applications  are  shown 
in  other  parts  of  this  section, 

14.  PULSE  TIMING  CIRCUITS 

Various  timing  problems  arise  in  pulse  systems.  Pulse  repetition  rate,  duration,  and 
delay  relative  to  a  reference  pulse  require  measurement  and  control. 

Voltage  wave  forms  frequently  used  for  pulse  timing  are  the  sinusoidal,  the  exponential, 
and  the  rectangular.  The  three  wave  forms  may  be  compared  for  basic  stability  resulting 
from  their  use. 

With  a  sinusoid,  if  the  timing  action  occurs  as  the  sinusoid  passes  through  the  value 
zero,  the  timing  is  not  greatly  affected  by  the  amplitude  of  the  sinusoid,  and  stability  of 
the  same  order  as  that  of  the  inductance  and  capacitance  is  obtained. 

In  commonly  used  exponential  timing  circuits,  the  timing  action  occurs  as  the  variable 
(voltage  or  current)  reaches  a  predetermined  amplitude.  The  timing  is  therefore  a  func- 
tion of  the  initial  amplitude  as  well  as  the  time  constant  of  the  exponential  change.  Sta- 
bility of  the  same  order  as  that  of  the  passive  circuit  elements  requires  compensation  for 
changes  in  tube  characteristics  and  operating  voltages.  Stability  is  generally  improved 
by  using  only  the  early  and  most  rapidly  changing  part  of  the  exponential  wave.  Timing 
in  relaxation  oscillators  is  generally  done  by  exponential  changes  to  predetermined  ampli- 
tudes. 

A  rectangular  wave  sent  through  a  delay  line  yields  a  timing  wave  in  which  the  entire 
amplitude  change  occurs  at  the  most  useful  time.  The  stability  obtained  is  limited  by 
the  allowable  volume  and  complexity  of  the  line. 

Repetition  Rate.  A  conventional  sine-wave  oscillator  followed  by  a  relaxation  or  shap- 
ing circuit  may  be  used  to  obtain  synchronizing  pulses  having  a  stable  repetition  rate  and 
short  rise  time.  The  sine  wave  may  be  converted  to  a  square  wave  by  several  clipper 
stages  as  discussed  in  article  12.  The  square  wave  may  then  be  differentiated  to  obtain  a 
suitable  trigger  pulse. 

Having  a  sinusoidal  oscillator  witk  good  frequency  stability,  a  phase  shifter  followed  by 
shaping  circuits  may  be  used  to  obtain  a  second  pulse  with  accurately  controlled  delay 
relative  to  an  unshifted  trigger  pulse.  This  system  has  been  used  in  radar  equipment  to 
measure  range  accurately. 

Repetition  rate  may  be  determined  by  a  free-running  relaxation  oscillator  such  as  a 
multivibrator,  blocking  oscillator,  or  thyratron.  The  repetition  rate  is  then  usually  more 
subject  to  drift  than  if  an  L-C  oscillator  is  used,  but  a  suitable  trigger  pulse  and  a  large 
range  of  repetition  rates  are  economically  obtained.  Figure  13  shows  a  relaxation  oscillator 


lutout         n 

fl 


FIG.  13.    Repetition  Rate  Determined  by  an  Exponential  Wave  Form 

providing  trigger  pulses  of  fixed  duration  as  the  repetition  rate  is  changed.  The  pulse 
duration  is  determined  by  the  exponential  charging  of  capacitor  C  through  r%  and  the  low 
grid-to-cathode  resistance  of  tube  T-l.  After  each  pulse  the  grid  of  tube  T-l  is  driven 
negative  for  a  period  determined  by  the  exponential  discharging  of  capacitor  C  through  TI 
and  the  relatively  small  resistance  of  r2  and  tube  T-2.  The  period  between  pulses  is  approx- 
imately proportional  to  the  resistance  n. 

In  some  applications  a  delay  line  may  be  used  to  control  repetition  rate  and  simultane- 
ously supply  synchronizing  pulses  at  desired  times  in  each  period.    Thus,  in  Fig.  14,  the 


9-20 


PULSE  TECHNIQUES 


blocking  oscillator  generates  pulses  which  after  traversing  the  delay  line  are  applied  to  the 
trigger  tube  and  initiate  new  pulses. 

+B  abed 

;  I 


_5 


Ht 

rt  1  5-  -._  L. 

Delay  =  £;2            o 

ir= 

tf_  .                       ! 
-c 

r — ^-Trigger  tube 
[locking  oscillator 
Putse       la 


-0 


* i 

td — 4  ^Trigger  for  pulse  2o 

FIG,  14.    Repetition  Rate  Determined  by  a  Rectangular  Wave  Form 

Btiratioiu  Pulses  having  a  required  duration  may  be  generated  by  relaxation  circuits 
or  by  pulse-forming  lines  and  thyratrons,  as  discussed  in  article  15,  "Pulse  Modulation 
of  an  Oscillator."  Alternatively,  sinusoidal  or  other  wave  forms  may  be  passed  through 
pulse  shaping  circuits  (see  above)  to  produce  pulses  of  the  required  duration. 

4-B 


Output 


Toward  +  B 


~LT 


Sync. 


FIG.  15.    Poise  Duration  Determined  by  an  Exponential  Wave  Form 

A  cathode-coupled  one-pulse  relaxation  circuit  is  shown  in  Fig.  15  to  illustrate  control 
of  pulse  duration  by  exponential  voltage  change. 

Blocking  oscillators  deliver  maximum  power  during  the  pulse  and  therefore  are  adapted 
for  high-power  pulse  equipment.  Nominal  control  of  pulse  duration  is  obtained  by  means 


-fB 


-c     •=• 

False  Diu-atkm  Determined  by  a  Rectangular  Wave  Form 


of  grid  capacitors  and  con- 
trol of  transformer  induct- 
ance, in  the  circuit  of  Fig, 
12.  Precise  duration  and 
better  pulse  shape  are  ob- 
tained by  means  of  a  delay 
line  as  in  Fig.  16.  The 
transformer  inductance  is 
made  large;  the  feedback, 
the  line  impedance,  and 
the  line  terminating  resist^ 
ance  are  so  chosen  that 
the  oscillator  is  shut  off 
when  the  wave  of  grid 


PULSE  MODULATION  OF  AN  OSCILLATOR 


9-21 


current  reflected  from  the  open  end  of  the  line  reaches  the  terminating  resistor.  The 
pulse  duration  is  then  determined  mainly  by  the  round  trip  delay  of  the  line  and  is  little 
affected  by  variations  in  other  circuit  elements. 

Delay.  A  complex  wave  may  be  delayed  by  passage  through  a  delay  line  while  main- 
taining substantially  the  same  wave  form.  If  it  is  only  required  that  a  delayed  pulse  be 
produced,  the  initiating  pulse  may  trigger  a  relaxation  oscillator,  and  the  trailing  edge  of 
the  relaxation  pulse  may  be  used  after  differentiation  as  the  delayed  pulse.  The  delay 
of  the  trailing  edge  of  the  relaxation  pulse  may  be  varied  in  accordance  with  a  desired 


Delay  line,  td,  ZQ 


+B 
(a)  Delay  Line 


ir 


Sync. 


(6)  Relaxation  Oscillator  (Optional  Pulse-time  Modulation) 


H-B 

(c)  Resonant  Circuit 
FIG.  17.    Circuits  for  Delaying  Pulses 

modulation  for  obtaining  pulse-time  modulation  in  a  communication  system.  Special 
cathode-ray  tubes  have  been  developed  to  facilitate  time  modulation  of  the  many  time- 
sharing pulse  channels  in  pulse-time  multiplex  systems.  In  another  method,  a  resonant 
circuit  is  keyed  on  or  suddenly  shocked,  and  a  delayed  pulse  is  produced  after  a  fraction 
of  a  cycle  of  oscillation.  Typical  circuits  using  (a)  "the  delay  line,  (6)  the  relaxation  oscil- 
lator, and  (c)  the  resonant  circuit  are  shown  in  Fig.  17. 

The  delay  line  can  accept  pulses  having  random  spacing  such  as  might  be  encountered 
in  the  output  of  a  receiver.  The  other  circuits  illustrated  must  pass  through  a  complete 
delay  and  recovery  cycle  between  pulses  and  therefore  are  best  used  with  isolated  pulses. 


15.  PULSE  MODULATION  OF  AN  OSCILLATOR 

An  oscillator  may  be  made  to  operate  in  pulses  by  including  in  its  grid  circuit  a  capacitor 
which  charges  during  the  oscillation  and  blocks  or  stops  the  oscillation.  This  is  one  form 
of  the  blocking  oscillator.  Oscillations  start  again  after  the  capacitor  has  discharged  suffi.- 


9-22 


PULSE  TECHNIQUES 


Output 


Bias 
FIG.  18.    Pulse-modulated  Carrier-frequency  Oscillator 


ciently.    The  pulse  recurrence  rate  and  the  pulse  duration  are  approximately  controlled 
by  the  time  constants  in  the  circuit,  but  only  with  limited  stability. 

The  recurrence  rate  of  a  blocking  oscillator  may  be  stabilized  by  the  use  of  synchronizing 
circuits  to  start  tube  conduction  sooner  after  each  pulse  than  would  otherwise  happen. 

The  pulse  duration  may 
be  stabilized  by  making 
a  time-delay  network 
shut  off  the  oscillator. 

The  oscillator  in  Fig. 
18  has  provision  for  syn- 
chronization by  a  posi- 
tive pulse  applied  to  the 
grid.  A  delay  line  in  the 
cathode  circuit  shuts  off 
the  oscillator  after  a 
time  interval  of  twice 
the  line  delay.  The  dis- 
charging resistor  r  is 
usually  many  times  as 
large  as  Z^  the  wave 
impedance-  of  the  line. 
At  the  start  of  the  pulse 
the  cathode  current  flows  through  an  impedance  approximately  equal  to  Z0,  and  develops 
a  voltage  wave  B  which  travels  through  the  line.  The  wave  is  reflected  with  the  same 
polarity  by  the  open  far  end  and  returns  to  increase  the  voltage  at  the  cathode  to  approxi- 
mately twice  the  initial  voltage,  stopping  the  oscillator. 

Pulse  modulation  may  also  be  accomplished  by  grid  or  plate  modulation  of  an  oscillator 
or  amplifier.  If  high  power  is  required,  plate  modulation  of  an  oscillator  is  generally  most 
economical.  With  plate 
modulation,  the  high  volt- 
age is  impressed]  on  the  os- 
cillator tube  only  during  the 
pulses.  The  design  of  the 
oscillator  tube  is  thereby 
made  easier,  or  conversely 
the  permissible  maximum 
plate  voltage  on  a  given 
tube  design  may  be  in-  Driver 

creased   far    beyond   the 
direct- voltage  rating. 

A  vacuum-tube  modulator  is  shown  in  Fig.  19.  The  modulator  tube,  cut  off  between 
pulses,  is  driven  hard  so  that  it  passes  high  current  through  low  impedance  during  the 
pulses.  The  modulator  acts  as  a  switch  connecting  the  load  to  the  B  supply  only  during 
the  pulses. 

A  thyratron  pulse  modulator  is  shown  in  Fig.  20.  The  artificial  line  is  charged  through  a 
high  resistance  rc  between  pulses.  By  the  use  of  the  line  instead  of  a  single  capacitor,  a 


-fB  -C 


Modulator 
FIG.  19.    Vacuum-tube  Modulator 


-  oj^iTY/iry^j^^YyT^o 


!' 

4-B 


Sync, 


— C 
FIG.  20.    Thyratron  Modulator 


rectangular  pulse  may  be  obtained  rather  than  an  exponentially  decaying  pulse.  The 
thyratron  discharges  the  line  into  the  load  and  must  deionize  before  the  line  can  recharge. 
3>eionization  is  expedited  by  making  the  thyratron  resistance  plus  the  transformed  load 
rcsisliuace  somewhat  less  than  the  line  impedance  so  that  the  thyratron  plate  is  driven 
sligfetJty  negative  just  after  the  pulse. 


MODULATING  THE    CHARACTERISTICS  OF  PULSES      9-23 


For  high-voltage  applications,  the  preferred  form  of  artificial  line  is  the  "Guillemin 
line,"  shown  in  Fig.  29  (e)  below.  Also,  the  thyratron  may  be  replaced  by  a  spark  gap. 
Higher  efficiency  and  faster  charging  of  the  line  may  be  obtained  by  replacing  the  charging 
resistor  rc  by  an  inductor.  The  line  is  then  charged  in  a  half-cycle  of  a  charging  oscillation, 
between  pulses,  to  approximately  twice  the  power  supply  voltage.  A  diode  may  be  con- 
nected in  series  with  the  inductor  to  prevent  discharge  back  into  the  power  supply. 

Pulse  modulation  imposes  on  an  oscillator  severe  requirements  of  fast  starting  and  stop- 
ping. The  oscillator  must  ordinarily  start  from  the  background  noise  in  the  circuit  and 
build  up  to  maximum  amplitude  in  a  time  interval  which  is  much  shorter  than  the  pulse 
duration.  Random  variations  in  the  noise  amplitude  cause  random  variations  in  starting 
time,  termed  "jitter." 

To  explain  the  starting  and  stopping  time,  the  essential  elements  of  the  oscillator  are 
indicated  in  Fig.  21.  They  are  the  resonator  CL  and  the  conductance  g.  During  the 
starting  time  of  a  pulse,  the  net  conductance  is  made  nega- 
tive by  feedback;  during  the  stopping  time  it  is  positive  as 
determined  by  the  damping  of  the  circuit  augmented  by 
its  useful  load. 

The  starting  or  stopping  time  constant  of  oscillations  is 
2C/g  The  starting  time,  for  the  oscillations  to  build  up 
by  regeneration  from  the  noise  level,  is  of  the  order  of  20 
times  the  time  constant  determined  by  the  negative  con- 
ductance, and  the  jitter  is  somewhat  less  than  this  time 
constant.  The  stopping  time  is  about  equal  to  the  stopping  time  constant,  since  the  oscil- 


I 
T 


Equivalent  Circuit  of  an 
Oscillator 


lation  is  merely  damped  from  its  peak  value  after  the  regeneration  is  cut  off. 
are  shown  in  Fig.  8,  p.  9-06. 


These  effects 


16.  MODULATING  THE  CHARACTERISTICS  OF  PULSES 

Preceding  sections  of  this  chapter  have  indicated  ways  of  forming  pulses  and  controlling 
their  characteristics.  Coding  of  pulses  for  identification  of  the  source  or  for  transmission 
of  information  may  be  accomplished  by  mechanical  switching  or  by  electronic  switching 
or  modulation.  Pulse  duration  may  be  changed  by  connecting  or  disconnecting  sections 
in  a  delay  line  used  to  control  a  blocking  oscillator.  Similarly,  the  constants  of  a  shaping 
circuit  or  relaxation  circuit  may  be  switched. 

The  circuit  shown  in  Fig.  17(6)  permits  electronic  modulation  of  pulse  width  such  as 
might  be  used  in  a  communication  system.  Differentiation  of  the  output  pulse,  as  shown 
in  the  figure,  yields  a  pulse  having  electronically  modulated  phase  relative  to  a  base  or 
marker  pulse.  Pulse  phase  or  time  may  also  be  modulated  at  an  audio  rate  by  addition 
of  audio  voltage  to  linear  sawtooth  timing  pulses  as  shown  in  Fig.  22.  No  plate  current 


•  Delay  of  modulated  pulse 
FIG.  22.    Pulse-time  Modulator 

flows  until  the  sum  of  the  audio,  the  sawtooth,  and  the  bias  voltages  exceeds  the  cutoff 
voltage  of  the  mixer  tube.  The  timing  of  the  leading  edge  of  the  resulting  plate-current 
pulse  varies  in  direct  proportion  to  the  instantaneous  value  of  the  audio  voltage.  The  mixer 


9-24 


PULSE  TECHNIQUES 


output  pulse  may  be  differentiated  and  shaped  to  obtain  the  desired  wave  form  for  pulse 
modulation  of  a  carrier  wave. 

A  group  of  pulses  may  be  obtained  by  shaping  a  sine  wave  which  has  the  proper  perio- 
dicity, to  obtain  trigger  pulses  for  a  relaxation  circuit.  The  sine  wave  is  initiated  and 
ended  by  a  gate  pulse  long  enough  to  permit  generation  of  the  required  number  of  pulses. 
Figure  23  shows  one  such  arrangement.  Triode  A  is  conducting  normally.  A  negative 


FIG.  23.    Multiple-pulse  Generator 

gate  pulse  cuts  off  the  triode,  developing  a  positive  gate  and  a  free  oscillation  to  drive  the 
grid  of  the  pentode  J?.  At  the  end  of  the  gate  pulse  the  oscillations  are  quiokly  damped 
by  the  plate  resistance  of  the  triode.  The  number  of  pulses  may  be  changed  by  switching 
the  gate  pulse  duration. 

Figure  24  shows  the  elements  of  a  two-pulse  generator  using  a  delay  line.  A  modification 
of  this  arrangement  in  which  the  pulse  spacing  would  correspond  to  altitude  has  been 
proposed  for  use  in  air  navigation  (Lanac).  An  initiating  pulse  passes  through  a  delay 


ri 


FIG.  24.    Double-pulse  Generator 

line  and  is  absorbed  by  the  resistor  terminating  the  line.  The  line  is  tapped  to  permit 
application  of  the  pulse,  after  the  required  delay,  to  the  grid  of  a  tube.  The  direct  and 
delayed  pulses  are  mixed  to  produce  a  pair  of  pulses  with  variable  spacing. 


17.  PULSE  DETECTORS 

Pulses  are  distinguished  from  sinusoidal  and  steady  waves  by  rapid  changes  in  ampli- 
tude, large  ratio  of  peak  to  average  value,  small  duty  cycle,  and  wide  frequency  spectrum. 
Tfee  detection  of  pulses  requires  detecting  means  responsive  to  the  special  characteristics 
®l  praises,  tsaaially  to  t&mr  tisae  boundaries  as  distinguished  from  their  peak  amplitudes. 

Bandwidth  requirements  have  been  discussed  earlier  in  this  section.  Pulse  amplifiers 
»ay  isielml©  Sites  or  coupling  systems  which  favor  pulses  having  certain  characteristics, 


9-26 


PULSE  TECHNIQUES 


require  more  pulses  in  a  group.    The  timing  of  two  or  more  pulses  in  a  group  may  be  used 
for  altitude  coding  in  an  air  navigation  system  (Lanac)  as  mentioned  previously. 

Phase  or  time  modulation  of  pulses  in  a  communication  system  may  be  converted  to 
audio  signals  by  a  corresponding  type  of  detection.    Phase  modulation  may  be  detected  by 

providing  reference  pulses 
of  uniform  periodicity  and 
using  these  to  form  gate 
pulses  of  suitable  phase 
which  the  phase-modulated 
pulses  overlap  more  or  less 
in  accordance  with  the  mod- 
ulation. If  the  gate  and 
the  phase-modulated  pulses 
are  applied  to  a  coincidence 
mixer,  plate  current  will 
flow  during  the  interval  of 
pulse  overlap.  The  plate 
current,  averaged  over  a 
few  pulses,  is  then  modu- 
lated in  accordance  with 
the  phase  modulation  of 
the  pulses.  Special  beam- 
deflection  tubes  have  been 


Delayed- 


Mixer  output-^  |     [ 

FIG.  27.    Double-pulse  Decoder 


developed  for  phase  modulation  and  synchronous  detection  of  many  time-sharing  channels 
in  &  pulsed  multiplex  system. 

18.  VACUUM  TUBES 

Most  vacuum  tubes,  at  the  date  of  writing,  have  not  been  rated  for  pulse  applications 
(see  Section  4,  especially  pulse  tubes  such  as  magnetrons  and  thyratrons).  However, 
many  vacuum  tubes  have  been  used  in  pulse  applications  after  rough  estimation  of  their 
pulse  capabilities  from  their  ordinary  ratings.  The  principal  factors  are  peak  and  average 
cathode  current,  peak  and  average  electrode  voltages,  electrode  power  dissipations,  and 
tube  life. 

If  a  tube  has  class  C  amplifier  ratings,  the  cathode  current  for  long  pulses  may  safely 
be  4  or  5  times  the  class  C  average-current  rating.  The  average  current  should  not  exceed 
the  class  C  average-current  rating.  For  pulses  a  few  microseconds  in  duration,  peak  cur- 
rents of  about  50  times  the  class  C  average-current  rating  may  be  taken  from  oxide-coated 
cathodes,  with  tube  life  commonly  exceeding  500  hours.  Pulse  current  densities  of  10  to 
20  amp  per  sq  cm  are  used  with  specially  processed  oxide-coated  cathodes.  The  cathodes 
must  be  heated  to  their  operating  temperature  before  large  currents  are  permitted. 

The  ordinary  peak-voltage  ratings  of  tubes  may  often  be  safely  exceeded  in  pulse  appli- 
cations. Insulation  breakdown  inside  the  tube,  electrolysis  of  the  glass,  or  poor  plate- 
current  cutoff  characteristics  may  fix  the  permissible  peak  voltage.  The  peak  plate  voltage 
in  class  C  applications  is  about  twice  the  average  plate  voltage;  this  is  usually  a  justifica- 
tion for  permitting  pulse  peak  voltages  twice  the  rated  average  plate  voltage.  If  the  plate 
voltage  is  applied  only  during  pulses,  the  sparkover  voltage  may  set  the  limit  at  a  much 
higher  voltage. 

Beam  power  tubes  in  pulse  modulators  are  operated  at  high  screen  voltage  in  order  to 
reduce  the  driving  power  required.  The  plate  voltage  is  usually  low  during  pulses  of 
current  flow,  so  that  average  plate  power  dissipation  is  low.  Screen  dissipation  is  high, 
however,  and  often  limits  average  power.  Permissible  limits  of  plate  and  screen  dissipa- 
tion may  be  reduced  as  peak-voltage  ratings  are  increased. 

The  space-charge  effects  which  suppress  secondary  emission  from  the  plate  in  beam 
tubes  also  tend  to  limit  the  flow  of  pulse  current  at  low  plate  voltage.  The  usually  desired 
properties  of  beam  tubes  are  even  more  important  for  pulse  amplifiers  of  very  high  power 
used  as  modulators.  Secondary  emission  from  the  control  grid  is  minimized  to  avoid  loss 
of  control  caused  by  reverse  grid  current  and  to  prevent  parasitic  oscillation. 

A  number  of  thyratrons  specifically  intended  for  pulse  modulation  are  available.  Minia- 
ture types  deliver  pulses  of  peak  power  of  kilowatts;  larger  types  deliver  megawatts.  Fast 
dekmization  is  obtained  in  tubes  filled  with  the  lighter  gases  such  as  hydrogen  and  helium, 
slower  deionization  with  argon  and  xenon.  Cathode  current  densities  are  of  the  same  order 
as  m  higfe-vactium  tubes  but  with  much  less  voltage  drop  between  plate  and  cathode.  The 
internal  voltage  drop  is  high  at  the  start  of  ionization,  very  rapidly  decreasing  as  ionization 
rme&es  saturation.  A  maximum  rate  of  rise  of  current  is  specified  to  limit  the  plate  dissi- 


PULSE  TRANSFORMERS 


9-27 


pation  and  the  ion  bombardment  of  the  cathode  at  the  start  of  the  pulse.  The  rate  of  rise 
of  current  is  usually  limited  by  some  inductance  in  series  with  the  plate  of  the  thyratron. 
It  is  worthy  of  mention  that,  at  the  high  currents  used  in  pulse  work,  the  transconduct- 
ance  of  vacuum  tubes  is  greater  than  usual,  and  transit  time  is  reduced  by  the  correspond- 
ing high  voltages.  The  performance  of  a  tube  in  a  pulsed  oscillator  is  therefore  often  better 
than  its  continuous-wave  performance  at  high  carrier  frequencies  subject  to  transit-time 
effects.  In  fact,  some  types  will  oscillate  efficiently  in  high-power  pulses  at  frequencies 
so  high  that  continuous  oscillation  is  impossible  at  their  Tnq.xJTn  \iirt  d-c  ratings. 

19.  PULSE  TRANSFORMERS 

The  problem  of  designing  a  pulse  transformer  is  generally  similar  to  that  of  designing 
an  audio-frequency  transformer  but  for  higher-frequency  components  with  corresponding 


-f-B  -C 
(a)  Interstage  Transformer 


(6)   Equivalent  Circuit  of  1:1  Transformer 


Rise  and  ripple  of 
due  to  Lj,  Clt  C2,  C3 


Decay  of  £2 
due  to  Tlt  T2,  \- 


Backswlng  of  £2 
duetoLm;  rlt 
Ci,  C2  *  A-T^and  r2  large 


(c)   Pulse  Transformer  Wave  Forms 
FIG.  28.    Pulse  Transformers 

reduction  in  size.  Maximum  mutual  inductance,  minimum  leakage  inductance,  and  min- 
imum incidental  capacitance  are  desired.  The  ratio  of  maximum  to  minimum  essential 
frequency  components  involved  in  faithfully  transforming  a  single  pulse  wave  form  may 
be  100  to  1.  If  the  pulse  duration  may  vary  by  10  to  1>  the  frequency  ratio  may  be  1000 
to  1.  The  requirements  for  high-power  modulator  transformers  are  usually  much  less 
severe  in  frequency  ratio. 

A  rough  estimate  of  transformer  specifications  may  be  obtained  by  analysis  of  the 
circuit  of  Fig.  28  (a)  and  the  equivalent  circuit  shown  in  Fig.  28(6).  A  turns  ratio  of  1  to 
1  is  assumed.  The  mutual  inductance  Lm  may  be  neglected  in  estimating  the  rise  time 


0-28  PULSE  TECHNIQUES 

of  the  pulse.    If  n  and  Ci  are  small,  the  rise  time  is  roughly  VLiCz  or  Li/rz,  whichever  is 
greater.    The  magnetizing  current  at  the  end  of  the  pulse  is 


if  the  pulse  duration  is  T*  The  decrease  in  voltage  during  the  pulse  and  the  backswing 
at  the  trailing  edge,  both  of  which  are  caused  by  the  magnetizing  current,  are  roughly  in 
the  ratio 

Affa_    T     nr2  (2) 

Et   =  Lm  TI  4-  rz 

if  ri  and  rz  are  small  and  linear.  If  n  and  r3  become  large  on  reversal  of  polarity,  the  back- 
swing  is  determined  by  Im,  Lm,  Ci  -f  £2,  Cs,  and  rm.  rm  represents  the  apparent  shunt 
resistance  of  eddy  currents  and  hysteresis  in  the  core  loss. 

To  reduce  eddy  currents  or  skin  effect  in  the  core,  very  thin  laminations  are  used  for 
pulse  transformers.  Permalloy  and  silicon-steel  tapes  from  0.0001  to  0.003  in.  in  thickness 
are  wound  into  toroidal  cores  which  are  sliced  for  assembly  with  coils.  Reduction  of  skirt 
effect  in  the  core  permits  greater  ratios  of  mutual  to  leakage  inductance  by  increasing  the 
depth  of  penetration  and  the  resulting  effective  permeability  of  the  core  volume.  The 
available  flux  swing  in  the  core  may  sometimes  be  increased  by  the  use  of  reverse- 
magnetizing  direct  current.  Because  pulse  wave  forms  are  usually  unidirectional,  and  the 
core  has  some  retentivity,  saturation  may  otherwise  occur  with  a  small  flux  swing,  per 
pulse.  Some  core  materials  especially  suited  for  pulse  applications  are  listed  below. 

Because  of  restrictions  on  leakage  inductance,  high-voltage  pulse  transformers  are 
difficult  to  insulate.  Impregnation  with  oil  or  polymerized  resins  to  eliminate  air  spaces 
and  moisture  are  effective  in  preventing  corOna  and  raising  the  breakdown  voltage. 

Table  1.    Core  Materials  for  Pulse  Applications 

Name  and  Manufacturer  Description 

Hypersil  Wound  silicon-steel  tape:  grain-oriented  0.002  in1 

Wesiinghouse  Electric  Corp.  thick  tape  for  high  quality;  plain  0.003  in. 

tape  for  less  critical  application. 

Permalloy  Wound  permalloy  tape:  from  0.0001  to  0.002  in. 

Bell  Telephone  Laboratories  thick;  several  different  alloys. 

Silicon  NicaJloy  Conventional  laminations:  type  B9W4A.    * 

General  Electric  Co. 

Sinimax  and  Monomax  Conventional     laminations:     several     different 

Allegheny  Ludlum  Steel  Corp.  alloys  (permalloys). 

20.  DELAY  LINES  * 

A  delay  line  is  a  network  used  for  storing  the  energy  of  a  pulse  pattern  and  delivering 
out  the  same  energy  at  a  later  time.  (See  Section  5;  Networks,  Lines,  Transients.)  The 
simplest  form  is  a  transmission  line,  in  which  case  the  delay  is  determined  by  the  distance 
along  the  line  at  a  velocity  somewhat  less  than  the  velocity  of  light.  While  the  transmission 
line  or  wave  guide  may  give  the  best  performance,  it  requires  a  length  too  great  for  most 
purposes,  so  more  concentrated  forms  have  been  devised  to  save  space. 

Delay  lines  are  used  for  two  general  purposes:  one  is  the  relative  timing  of  different 
operations;  the  other  is  the  delayed  reproduction  of  a  pulse  pattern.  The  former  may  be 
tolerant  of  distortion  in  the  delay  process  and  can  even  be  accomplished  by  other  kinds 
of  timing  circuits  such  as  relaxation  oscillators.  The  delayed  reproduction  of  a  pulse 
pattern,  however,  places  the  most  severe  requirements  on  a  delay  network,  especially 
against  amplitude  and  phase  distortion.  For  either  purpose,  precision  of  timing  may  be  a 
requisite  which  involves  directly  the  stability  of  the  network  and  indirectly  the  fidelity  of 
reproduction  and  the  stability  of  associated  circuits. 

Figure  29  shows  some  typical  concentrated  delay  networks.  They  rely  on  coils  for 
concentrating  the  inductance  and  on  dielectric  for  concentrating  the  capacitance.  Further 
concentration  of  inductance  may  be  obtained  by  iron-dust  cores  for  the  coils.  * 

The  continuous  coil  with  capacitive  loading  is  shown  in  Fig.  29  (a).  In  a  typical  form,  a 
coH  of  fine  enameled  wire  is  wound  on  a  flexible  core  of  insulation  material  having  such 
small  diameter  that  it  can  be  coiled.  The  winding  is  wrapped  with  thin  dielectric  tape  for 
eapadtive  loading,  and  the  other  conductor  is  provided  by  a  braided  sheath  of  fine 

*  Tfefe  article  by  Harold  A.  Wheeler. 


DELAY  LINES 


9-29 


enameled  wire.     The  sheath  acts  as  a  capacitive  shield  but  not  as  an  inductive  shield, 

because  the  latter  would  destroy  the  desired  inductance  of  the  coiled  inner  conductor. 

The  entire  line  is  protected  by  a  covering  of  insulation. 

In  a  continuous  coil,  the  mutual  inductance  along  its  length  has  the  effect  of  decreasing 

the  effective  inductance  at  higher  frequencies  relative  to  that  of  lower  frequencies,  which 

causes  phase  distortion.  In  Fig.  30,  the 
curved  dotted  line  (a)  shows  this  effect. 
Uniform  delay,  free  of  distortion,  requires 
a  straight  phase  curve  illustrated  by  the 
solid  line  (&).  It  has  been  found  possible 
to  approximate  this  ideal  by  introducing 
substantial  distributed  capacitance  in 
parallel  with  the  series  inductance,  as  is 
also  shown  in  Fig.  29 (a).  This  causes 
the  effective  inductance  to  increase  with 
frequency  and  can  be  designed  to  com- 
pensate for  the  opposite  effect  of  mutual 
inductance  along  the  coil.  Without  such 


-TTTT  T. 


Ll 
r  Total 

°J 


Continuous  Coif 


Total 


(6)  Continuous  Bifilar  Coil 


c/TTv^WrvTT^ 

-    T     T     T     T    .  C 


Lumped  Line 


oil        oU 

_Z2    .Z2 


(<i)  Open-circuit  Reflecting  Line  by 
Parallel  Components 

6/!  4f,  2f, 

r^S  r^S  r^S 

HL,JLiJLJlh 

ire;      'ic       lie     c 
2        T        2" 

o 1 


Frequency 
FIG.  30.    Phase  Distortion  of  Continuous  Coil 

compensation,  the  effect  may  be  reduced 
by  using  a  coil  diameter  very  small  as 
compared  with  its  length,  which  tends, 
however,  to  defeat  the  aim  of  concentra- 
tion. 

The  low-frequency  delay  (initial  phase 
slope)  of  a  delay  line  is 


••  VOL 


(3) 


in  terms  of  the  total  shunt  capacitance  C 
and  total  series  inductance  L  (including 
mutual  inductance)  .  The  corresponding 
wave  impedance  (characteristic  imped- 
ance or  image  impedance)  is 


ZQ 


(4) 


|  Open-circuit  Reflecting  Line  by 
Series  Components 

Any  of  the  following  sets  of  units  may 
be  specified,  the  last  (long  used  by  Dr. 
Alan  Hazeltine)  being  generally  most  convenient: 


FIG.  29.    Delay  Lines 


T 
C 
L 


seconds 
farads 
henrys 
ohms 


ohms 


mh 
kilohms 


As  an  example,  a  rather  long  coiled  line  of  1  mjuf  and  1  mh  has  a  delay  of  1  /is  and  a  wave 
impedance  of  1  kilohm. 

Another  form  of  coiled  line,  shown  in  Fig.  29(6),  is  made  of  two  coils  wound  in  opposite 
screw-^direotions  on  the  same  core.  One  coil  is  wound  in  one  direction  on  the  core,  then 
the  other  is  wound  on  top  in  the  opposite  direction.  The  insulation  may  be  only  the 
enamel  on  the  wires,  or  it  may  also  include  some  thin  layers  of  dielectric.  This  method 
gives  the  inductance  of  a  two-layer  coil  with  convenient  capacitive  loading.  It  is  useful 
as  a  balanced  four-terminal  circuit  or  as  a  two-terminal  reflecting  line  with  far  end  on  open 
circuit  or  short  circuit. 

The  low-pass  filter  of  Fig.  29  (c)  is  a  lumped  delay  line.  Its  useful  bandwidth  is  some- 
what less  than  its  cutoff  frequency  and  decreases  further  with  more  sections.  The  con- 


9-30 


PULSE  TECHNIQUES 


stant-fe  type  (without  mutual  inductance)  has  concave  phase  distortion  as  shown  in  Pig. 
31  (a).  By  converting  to  an  m-derived  type,  by  the  addition  of  mutual  inductance  M 
between  adjacent  inductors,  it  is  possible  to  approach  the  ideal  linear  phase  (5)  to  the 
-  approximation  indicated  by  curve  (c).  For  a  line  of  many 
sections,  the  optimum  value  of  the  design  parameter  m  is 
V3/2  =  1.22.  The  number  of  sections  required  is 


N  =  -fJT 

m 


(5) 


0  Frequency  fa 

FIG.  31.    Phase  Distortion  of 
Lumped  Line 


in  terms  of  the  cutoff  frequency  Je  (Me)  and  the  delay  T  (jus). 
As  an  example,  a  pulse  1  fts  wide  can  be  delayed  2  jus  by  a 
network  of  fe  =  I  Me  and  N  «  5  sections. 

In  a  concentrated  delay  line  with  phase  correction,  espe- 
cially if  the  delay  is  many  times  the  permissible  widening  of  the 
pulse,  the  attenuation  increasing  with  frequency  is  likely  to 
impose  the  practical  limitation  on  the  useful  bandwidth.  Therefore  all  losses  must  be 
minimized  to  secure  the  optimum  design  in  limited  space. 

A  reflecting  line  is  often  useful  and  has  the  advantage  of  a  round  trip  which  doubles  the 
delay.  The  voltage  polarity  of  reflection  is  direct  or  reverse,  depending  on  whether  the 
far  end  of  the  line  is  an  open  or  short  circuit.  Ordinarily  the  near  end  is  terminated  with  a 
resistance  load  matching  the  line  impedance  to  preclude  multiple  reflection. 

A  reflecting  line  becomes  a  two-terminal  network  whose  essential  properties  are  expressed 
by  the  variation  of  impedance  with  fre- 
quency. An  ideal  delay  line  of  uniform 
delay  and  no  losses  has  a  pure  react- 
ance as  shown  in  Fig.  32  if  the  far  end 
is  on  open  circuit.  The  alternate  zeros 
and  poles  are  at  odd  and  even  multiples 
of  the  fundamental  frequency: 


(6)       - 


At    this   frequency,    the  length  is  1/4       FIG.  32.    Impedance  of  Open-circuit  Reflecting  Line 
wavelength  in  the  line. 

The  impedance  of  a  reflecting  line  can  be  duplicated  over  a  limited  frequency  bandwidth 
by  other  two-terminal  impedance  networks  which  sometimes  have  advantages.  Figure 
29  (d)  presents  the  reactance  pattern  of  Fig.  32  by  a  parallel  connection  of  several  series- 
tuned  circuits  having  equal  values  of  inductance  but 
such  values  of  capacitance  as  to  resonate  at  odd 
multiples  of  the  fundamental  frequency.  Figure 
29  (e)  presents  the  same  reactance  pattern  by  a  series 
connection  of  parallel-resonant  circuits  having  equal 
values  of  capacitance  and  resonated  at  even  multi- 
ples. This  last  network  has  a  great  advantage  in 
pulse  discharging  functions  (as  proposed  by  E.  A. 
Guillemin)  because  only  the  isolated  capacitor  C  has 
to  take  the  high-voltage  charge  preparatory  to  the 
discharge;  the  other  capacitors  can  have  much  lower 
voltage  ratings. 

When  a  charged  delay  line  is  discharged  into  a 

matched  load,  the  output  is  a  pulse  whose  width  is       (&)  Concave  Phase  Curvature  (Fig.  31) 
the  round-trip  delay  2TT  as  shown  in  Fig.  33.   The 
pulse  starts  suddenly  by  the  discharge  of  energy 


R 


—Time 


(Or)  Convex  Phase  Curvature  (Fig,  30) 


-Time 


2T 


-Time 


(c)  Phase  Correction 


FIG.  33.    Pulse  Discharge  of  a  TJ"e  into  a 
Matched  Load 


FIG.  34.    Delay  of  a  Pulse  by  a  Reflecting 

Line  on  Open  Circuit,  Showing  the  Effects 

of  Distortion 

near  the  terminals,  but  the  trailing  edge  is  sloping  because  the  delayed  energy  is  subiect 
to  *b©  distortion  in  the  line. 


Time 


BIBLIOGRAPHY  9-31 

Figure  34  shows  the  distortion  of  a  pulse  subjected  to  round-trip  delay  in  a  reflecting 
line  on  open  circuit.  In  every  case  the  pulse  is  decreased  in  area  by  the  attenuation  and 
is  widened  by  the  amplitude  distortion  limiting  the  frequency  bandwidth.  The  effects 
of  phase  distortion  can  be  estimated  by  the  method  of  paired  echoes.  The  leading  tran- 
sient (a)  and  the  trailing  transient  (6)  are  charac- 
teristic of  convex  and  concave  phase  curvature, 
shown  in  Figs.  30  and  31.  The  former  occurs  in  con- 
tinuous coiled  lines  and  the  latter  in  lumped  lines.  In 
either  case,  phase  correction  (Fig.  34c)  removes  the 
transient  oscillation,  since  the  oscillation  is  caused  _ 

bv  subnormal  or  abnormal  delay  of  the   higher-     -n^     oe     T\  i          JT>  •,    *     -n  , 

u<y  ,       r  •     XT.          i  Fis* 35-    Delay  and  Reversal  of  a  Pulse 

frequency  components  of  energy  in  the  pulse.  by  a  Reflecting  Line  on  Short  Circuit 

If  the  direct  and   reflected   pulses  have  to  be 

separated,  a  delay  line  on  short  circuit  (or  an  equivalent  two-terminal  network)  is  used  to 
secure  the  result  shown  in  Fig.  35.  Since  the  respective  pulses  are  of  opposite  polarity, 
one  or  the  other  can  be  selected  by  a  rectifier. 

BIBLIOGRAPHY- 

Note.  Under  each  heading,  the  references  are  listed  chronologically  (except  for  the  *TB"  items, 
whose  dates  of  publication  do  not  appear  in  their  titles).  The  "PB"  items  are  issued  by  Office  of 
Technical  Services,  U.  S.  Department  of  Commerce,  Washington  25,  D.  C. 

Asterisks  (*)  denote  some  of  the  principal  references  or  further  bibliographies  on  each  subject. 

Radar 

*  1.  Navy  Dept.,  Radar  System  Fundamentals,  Navships  900,017  April  1944. 

*  2.  Navy  Dept.,  Radar  Electronic  Fundamentals,  Navships  900,016,  June  1944. 

3.  D.  G.  Fink,  The  Radar  Equation,  Electronics,  Vol.  IS,  No.  4,  92-94  (April  1945). 

4.  W.  E.  Moulic,  Operational  Elements  of  a  Radar  System,  Electronic  Industries,  Vol.  4,  No.  5, 

76-80,  225,  226  (May  1945). 

5.  Jordan  McQuay,   Practical  Radar,   Radio  News:  Part  I,   Vol.  33,  No.  6,  29    (June   1945); 

Part  2,  Vol.  34,  No.  1,  38  (July  1945) ;  Part  3,  VoL  34,  No.  2,  39  (August  1945) ;  Part  4, 
Vol.  34,  No.  3,  40  (September  1945);  Part  5,  Vol.  34,  No.  4,  44  (October  1945). 

*  6.  R.  B.  Colton,  Radar  in  the  United  States  Army,  Proc.  I.R.E.,  VoL  33, 740-753  (November  1945). 
7.  Report  on  Wartime  Electronic  Developments,  Electronics,  Vol.  18,  No.  11,  92-93  (November 

1945). 
*8.  Radar  Specifications,  Electronics,  Vol.  18,  No.  11,  116-119  (November  1945). 

*  9.  The  SCR-584  Radar,  Electronics:  Part  I,  VoL  18,  No.  11,  104-109  (November  1945) ;  Part  II, 

Vol.  18,  No.  12,  104-109  (December  1945). 

10.  C.  W.  Watson,  Ground-controlled  Approach  for  Aircraft,  Electronics,  Vol.  18,  No.  11,  112-115 

(November  1945). 

11.  H.  A.  Straus,  L.  J.  Rueger,  C.  A.  Wert,  S.  J.  Reisman,  M.  Taylor,  R.  J.  Davis,  and  J.  H. 

Taylor,  The  MPG-1  Radar,  Wectronics:  Part  I,  Vol.  18,  No.  12,  92-97  (December  1945); 
Part  II,  Vol.  19,  No.  11,  110-117  (January  1946). 

12.  W.  C.  Hendricks,  Lightweight  Radar  for  Early  Warning,  Communications,  Vol.  26,  54  (January 

1946). 

*  13.  W.  C.  Tinus  and  W.  H.  C.  Higgins,  Early  Fire-control  Radars  for  Naval  Vessels,  Bell  Sys. 

Tech.  J.t  VoL  25,  1-47  (January  1946). 

*  14.  H.  A.  Zahl  and  J.  W.  Marchetti,  Radar  on  50  Centimeters,  Electronics,  VoL  19:  Part  1,  No.  1, 

98-104  (January  1946);  Part  2,  No.  2,  98-103  (February  1946). 

15.  W.  T.  Spicer,  The  GCA  Landing  System,  Bendix  Radio  Engineer,  January  1946.    (Radar  on 

3000  Me  or  10,000  Me  with  PPI  display.) 

16.  H.  V.  Hermansen,  Bendix  Radio  Engineer,  January  1946. 

17.  A  Tool  for  Traffic  Control — Telerad,  Air  Transport,  VoL  4  /No.  1,  73-78  (January  1946) 

18.  I.  F.  Byrnes,  Merchant  Marine  Radar,  fl.C.A.  Rev.,  VoL  7,  54-66  (March  1946). 

19.  Jack  Mofenson,  Radar  Echoes  from  the  Moon,  Electronics,  Vol.  19,  No.  4,  92-98  (April  1946). 

*  20.  D.  A.  Quarles,  Radar  Systems  Considerations,  Elec.  Eng.,  Vol.  65,  Trans.,  pp.  209-215  (April 

1946). 

*  21.  G.  V.  Holdam,  S.  McGrath,  and  A.  D.  Cole,  Radar  for  Blind  Bombing,  Electronics,  VoL  19: 

Part  I,  No.  5,  138-143  (May  1946) ;  Part  II,  No.  6,  142-149  (June  1946). 

*  22.  O.  P.  Ferrell,  London  Radiolocation  Convention,  Radio,  Vol.  30,  No.  5,  26, 28,47, 48  (May  1946). 

23.  L.  N.  Ridenour,  Radar  in  War  and  Peace,  Elec.  Eng.,  VoL  65,  202-207  (May  1946). 

24.  R.  C.  Jensen  and  R.  A.  Arnett,  Air-borne  Radar  for  Navigation  and  Obstacle  Detection,  Elec. 

Eng.,  VoL  65,  No.  5,  307-313  (May  1946). 

25.  H.  Busignies,  P.  R.  Adams,  and  R.  I.  Colin,  Aerial  Navigation  and  Traffic  Control  with 

Navaglobe,  Navar,  Navaglide  and  Navascreen,  Elec.  Communication,  VoL  23,  No.  2,  113— 
143  (June  1946). 

26.  E.  I.  Green,  H.  J.  Fisher,  and  J.  G.  Ferguson,  Techniques  and  Facilities  for  Microwave  Radar 

Testing,  Bell  Sys.  Tech.  J.,  VoL  25,  No.  3,  435-482  (July  1946).  *<>• 

27.  Thomas  Grover  and  E.  C.  Kluender,  The  Electronic  Navigator,  Communications,  VoL  26,  No.  8, 

30,  36-39  (August  1946). 

*  28.  E.  G.  Schneider,  Radar,  Proc.  I.R.E.,  Vol.  34,  52&-57S  (August  1946). 

29.  L.  R.  Quarles  and  W.  M.  Breazeale,  Factors  Affecting  the  Range  of  Radar  Sets,  Elec.  Eng., 
VoL  65,  Trans.,  pp.  546-548  (August-September  1946). 

*  30.  L.  V,  Berkner,  Naval  Airborne  Radar,  Proc.  I.R.E.,  VoL  34,  671-706  (September  1946). 

31.  E.  D.  Hart,  Navigational  Rads-r  in  Merchant  Ships,  Electronic  Eng.,  VoL  18, 265-267  (September 

1946). 

32.  J.  H.  Cook,  Airborne  Search  Radar,  Bell  Lab.  Rec.,  VoL  24,  321-325  (September  1946). 


9-32  PULSE  TECHNIQUES 

33.  M.  L.  Lawrence,  A  100-kw  Portable  Radar  Transmitter,  Communications,  September  1946, 

34.  C.PB.*Barnel  Radar  Carrier-based  Planes,  Electronics,  Vol.  19,  No.  10,  100-105  (October  1946). 
35    H  B   Brooks  Weather  Forecasting,  Electronics,  Vol.  19,  No.  10,  84-87  (October  1946). 

36.  M.I.T.  Radar  School  Staff,  Principle*  of  Radar,  2nd  Ed.    McGraw-Hill  (1946). 
37    L.  H.  Lynn  and  O.  H.  Winn,  "Marine  Radar  for  Peacetime  Use,"  Elec.  Eng.t  Transactions, 
Vol.  65,  271  (May  1946).  < 

38.  PB1SQ42,  Navy  Dept.,  Radar  SU-1  with  PPI  display. 

39.  PB  19248,  War  Dept.,  Radar  APS-4A  on  9400  Me. 

40.  PB20810,  Navy  Dept.,  Radar  APS-4. 

41.  PB20S12,  War  Dept.,  Radar  APS-15  training  course. 

42.  PB20S13,  Navy  Dept.,  Radar  APS-3,  on  9400  Me. 

43.  PB2240Q,  Navy  Dept.,  Radar  CXAM-1  on  195  Me. 

44.  PB23435,  War  Dept.,  Radar  TPS-3. 

45.  PB23438,  War  Dept.,  Radar  SCR-6S2-A. 

46.  PB23447,  War  Dept.,  Radar  SCR-682-A. 

47.  PB24913,  Navy  Dept.,  Radar  SOS. 

48.  PB24933,  Navy  Dept.,  Radar  SR  with  A-type  and  PPI  displays. 

49.  PB24936,  Navy  Dept.,  Radar  SJ-1  with  PPI  display. 

50.  PB24938,  Navy  Dept.,  Radar  SA-3  on  200  Mo. 

Radio  Pulse  Altimeters 

61.  Heinrich  Lowry,  Electric  Proof  and  Measuring  of  the  Distance  of  Electrically  Conductive 

Bodies,  U.  S."  Pat.  1,585,591,  July  17,  1923-May  18,  1926. 

62.  Gregory  Breit  and  M.  A.  Tuve,  A  Test  of  the  Existence  of  the  Conducting  Layer,  Phys.  Rev., 

Vol.  28,  554-575  (September  1926). 

63.  R.  W.  Hart,  Measuring  Distance,  U.  S.  Pat.  1,924,156,  May  19,  1930-Aug.  29,  1933. 

64.  Ezekiel  Wolf,  Measuring  Distance,  U.  S.  Pat.  1,924,174,  May  19,  1930-Aug.  29,  1933. 

65.  D.  L.  Plaistowe,  Radio  Apparatus  for  Detecting  Aircraft,  U.  S.  Pat.  2,207,267,  July  11,  1938- 

July  9,  1940. 

66.  Joseph  Lyman,  Radio  Absolute  Altimeter,  U.  S.  Pat.  2,227,598,  July  3,  1937-  Jan.  7,  1941. 

*  67.  Albert  Goldman,  Pulse-type  Radio  Altimeter,  Electronics,  Vol.  19,  No.  6,  116-119  (June  1946). 
68.  P.  G.  Suizer,  Ionosphere  Measuring  Equipment,  Electronic*,  Vol.  19,  No.  7,  137-141  (July  1946). 

Navigation  and  Identification  by  Poises 

*  71.  Editor  (D.  G.  Fink),  The  Loran  System,  Part  I,  Electronics,  Vol.  18,  No.  11,  94-99  (November 

1945).      Loran   Receiver-indicator,   Vol.    18,    No.    12,    110-115    (December   1945).     Loran 
Transmitting  Stations,  Vol.  19,  No.  3,  109-115  (March  1946). 

72.  Editor,   Fundamentals   of  Radar  —  Pulse   Methods  Applied  to  Navigation,    Wireless    World, 

VoL  52,  23-26  (January  1946). 

73.  Editor,  Fundamentals  of  Radar.     5.  Beacons  Employing  Pulse  Technique,   Wireless  World, 

VoL  52,  55-56  (February  1946). 

74.  D.  Davidson,  Loran  Indicator  Circuit  Operation,  Elec.  Ind.,  Vol.  5,  No.  3,  84-93,  126,  128, 

130,  132  (March  1946). 

*  75.  S.  W.  Seeley,  Shoran  Precision  Radar,  Elec.  Eng.,  Vol.  65,  Trans.,  pp.  232-240  (April  1946). 

*  76.  J.  A.  Pierce,  An  Introduction  to  Loran,  Proc.  I.R.E.,  Vol.  34,  216-234  (May  1946). 
*77.  Hazeltine  Electronics  Corp.,  Lanac  Air  and  Marine  Navigation  (1946). 

78.  PB20811,  War.  Dept.,  Radar  Transponder  Beacon  APN-7  on  3300  Me. 

79.  PB23325,  P.  J.  Herbet,  Ultra  Portable  Racon, 
*80.  PB23434,  War.  Dept.,  Radar  RC-145-A. 

81.  PB23448,  War  Dept.,  Radar  RC-150B,  RC-151,  etc. 

82.  PB24920T  Navy  Dept.,  Radar  CPN-3. 

83.  PB24922,  Navy  Dept.,  Radar  BQ. 

}  Sonar 

*  91.  R.  J.  Evans,  Echo  Ranging  Sonar,  Electronics,  VoL  19,  No.  8>  88-93  (August  1946) 

*  92.  G.  B.  Shaw,  Echo  Depth  Sounder  for  Shallow  Water,  Electronics,  VoL  19,  No.  9,  88-92  (S*r>tem, 

ber  1946). 

93.  Thomas  Roddam,  Radar  in  Nature,  Wireless  World,  VoL  52,  286-288  (September  1946) 
©4.  PB19955,  Navy  Dept.,  Sonar  QBE,  ABE,  etc, 
95.  PB25497,  O.  R.  Smith,  Fish  Schools—  Location  by  Sonic  Equipment. 

Pulse  Techniques  and  Measurements 

*  101.  D.  G.  Fink,  ilfieriwcwe  Radar,  Vol.  1,  Theory  and  Practice  of  Pulsed  Circuits  (July  1942). 

*  102.  J.  G.  Brainerd,    Ultra-high-frequency  Techniques.     Van  Nostrand  (1942).     Chapter  4    triggei 

circuits,  pulse-sharpening  circuits  and  oscillators,  pp.  168-207. 

*  103.  Navy  Dept.,  Microwave  Techniques,  Navships  900028  (June  1944). 

104.  Aflfm  Easton,  Puke  Response  of  Diode  Voltmeters,  Electronics,  VoL  19,  No.  1,  146-149  (Jan- 

uary 1946). 
106.  Allan  Easton,  Measuring  Pufee  Cteacteristies,  Electronics,  Vol.  19,  No.  2,  150-154  (February 

*  106'  G  "0^  at  Frequeno  ies 


*  107-  '•  *1'-  iad  FacUities  for  Microwave  ^ 


*  108,  M.I.T,  Radar  School  Staff,  Principles  of  Radar,  2nd'  Ed.    McGraw-Hill  (1946). 

109.  PB21937,  War  Dept.,  Radar  Signal  Generator  TS-155B/UP  on  2700-2900  Me.* 

11®.  PB23432,  War.  Dept.»  Range  Calibrator  1-146. 

111.  PB23625,  R.  E.  Darrell,  Pulse  Voltmeter  Design  and  Analysis. 

112.  PB24921,  Navy  Dep*,,  PPI  repeater  VD. 

i?f"  £H!?SJ'  1*4  ^fe  S^ct^im .a.najy?er  for  Pulsed  oscillators  at  3000  Me. 

U4=.  FB32786^  Bnttou  Chance,  Precision  Time  Calibrator  and  Range  Measuring  System. 


BIBLIOGRAPHY  9-33 


Pulse  Tubes 

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122.  Editor,  Cavity  Magnetrons,  Electronics,  Vol.  19,  No.  1,  126-131  (January  1946). 

123.  H.  G.  Shea,  Theory  of  Magnetron  Tubes  and  Their  Uses,  Elec.  Ind.,  Vol.  5,  No.  1,  66-70 

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124.  J.  H.  Findlay,  Design  and  Construction  of  Radar  Series  Spark-gap  Tubes,  Elec.  Mfg.,  Vol.  37, 

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*  125.  J.  B.  Fisk,  H.  D.  Hagstrum,  and  P.  L.  Hartman,  The  Magnetron,  as  a  Generator  of  Centimeter 

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*  127.  R.  R.  Law,  D.  G.  Burnside,  R.  P.  Stone,  and  W.  B.  Whalley,  Development  of  Pulse  Triodes 

to  Give  1  Megawatt  at  600  Megacycles,  R.C.A..  Rev.,  Vol.  7,  253-264  (June  1946). 
128.  M.  Levy,  Power  Pulse  Generator,  Wireless  Eng.,  Vol.  23,  192-197  (July  1946). 

*  129.  Harold  Heinst  Hydrogen  Thyratrons,  Electronics,  Vol.  19,  No.  7,  96-102  (July  1946). 

130.  E.  A.  Coomes,  The  Pulsed  Properties  of  Oxide  Cathodes,  J.  Applied  Phys.,  Vol.  17,  647-657 


(August  1946). 
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131.  J.  J.  Glauber,  Radar  Vacuum-tube  Developments,  Elec.  Comm.,  Vol.  23,  306-319  (September 

1946). 

132.  Lloyd  P.  Hunter,  Energy  Build-up  in  Magnetrons,  J.  Applied  Phys.,  Vol.  17,  833-843  (October 

1946), 

133.  PB28647,  R.  C.  Fletcher  and  G.  M.  Lee,  Preliminary  Studies  of  Magnetron  Build-up. 

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*  142.  O.  S.  Puckle,  Time  Bases.    John  Wiley  (1943). 

143.  F.  E.  Terman,  Radio  Engineers1  Handbook,  pp.  511-516.    McGraw-Hill  (1943). 

*  144.  Navy  Dept.,  Timing  Circuits,  Navships  900,013  (May  1944). 

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147.  L.  B.  Tooley,  Gate  Circuits  for  Chronographs,  Electronics,  VoL  19,  No.  5,  144-145  (May  1946). 

148.  R,  Stang,  The  Blocking  Oscillator,  Radio  News,  VoL  36,  No.  3,  14-15,  20  (September  1946). 

I>elay  Lines 

151.  J.  P.  Blewett  and  others,  Delay  Lines.    General  Electric  Co.  reports  (1943). 

152.  M.  J.  DiToro,  Delay  Lines.    Hazeltine  Electronics  Corp.  reports  (1945). 

153.  M.  G.  E,  Golay,  The  Ideal  Low-pass  Filter  in  the  Form  of  a  Dispersionless  Lag  Line,  Proc. 

I.R.E.,  VoL  34,  138P-144P  (March  1946). 

154.  J.  M.  Lester,  Transient  Delay  Line,  Electronics,  VoL  19,  No.  4,  147-149  (April  1946). 

155.  H.  E.  Kallmann,  High-impedance  Cable,  Proc.  I.R.E.,  VoL  34,  348-351  (June  1946). 

156.  H.  E.  Kallmann,  Equalized  Delay  Lines,  Proc.  I.R.E.,  VoL  34,  646-657  (September  1946). 

157.  K.  H.  Zimmerman,  Spiral  Delay  Lines,  Elec.  Comm.,  VoL  23,  327-328  (September  1946). 

Pulse  Transformers 

161.  H.  A.  Wheeler,  Formulas  for  the  Skin  Effect,  Proc.  J.R.E.,  VoL  30,  412-424  (September  1942). 

162.  R.  Lee,  Iron-core  Components  in  Pulse  Amplifiers,  Electronics,  VoL  16,  No.  8,  115-117,  262-268 

(August  1943). 

*  163.  A.  G.  Ganz,  Permalloy  Tape  in  Wide-band  Telephone  and  Pulse  Transformers,  Elec.  Eng., 

VoL  65,  Trans.,  pp.  177-183  (April  1946). 
164.  PB32773,  J.  W.  Dunifon,  Pulse  Transformers. 

Computers 

171.  J.  T.  Potter,  A  Four-tube  Counter  Decade,  Electronics,  VoL  17,  No.  6,  110-113,  358,  360  (June 

1944). 

172.  Editor,  Preset  Interval  Timer,  Elec.  Ind.,  VoL  4,  No.  7,  97-99,  130,  134,  138, 142,  146  (July  1945). 

173.  V.  H.  Regener,  Directly  Coupled  Pentode  Trigger  Pairs,  Ret).  Sci.  Instr.,  VoL  17,  180-184  (May 

1946).    Decade  Counting  Circuits,  185-189. 

174.  A.  W.  Burke,  Super  Electronic  Computing  Machine,  Elec.  Ind.,  VoL  5,  No.  7,  62-67,  96  (July 

1946). 

175.  H.  G.  Shea,  Electronic  True  Decade  Counters,  Elec.  Ind.,  VoL  5,  No.  9,  82-84,  136  (September 

1946). 

176.  A.  Kip,  A.  Bousquet,  R.  Evans,  and  W.  Tuttle,  Design  and  Operation  of  an  Improved  Counting 

Rater  Meter,  Ret.  Sci.  Instr.,  VoL  17,  323-333  (September  1946). 

177.  B.  E.  Watt,  Current  Integrator,  Rev.  Sci.  Instr.,  VoL  17,  334-338  (September  1946). 

178.  A.  G.  Bousquet,  Radioactivity  Meter  for  Nuclear  Research,  Elec.  Ind.,  VoL  5,  No.  9,  88-89 

(September  1946). 

179.  I.  E.  Grosdoff,  Electronic  Counters,  R.C.A.  Rev.,  VoL  7,  438-447  (September  1946). 

180.  V.  H.  Regener,  Reversible  Decade  Counting  Circuit,  Rev.  Sci.  Instr.,  VoL  17,  375-376  (October 

1946). 

*  181.  R.  R.  Batcher  and  William  Moulic,    The  Electronic  Control  Handbook.     CaldweU-Clements 

(1946).    Section  3,  Chapter  3,  Counting  and  Timing  Controls,  pp.  178-195,  338. 
182.  S.  A.  Karff,  Electron  and  Nuclear  Counters.    Van  Nostrand  (1946). 

Picture  Transmission 

*  191.  RCA,  Television,  VoL  1  (July  1936);  Vol.  2  (October  1937). 

*  192.  J.  C.  Wilson,  Television  Engineering.    Sir  Isaac  Pitman  &  Sons,  London  (1937).    Chapter  12, 

Physical  Limitations,  pp.  420-437. 


9-34  PULSE  TECHNIQUES 

193.  J.  C.  Wilson,  Channel  Width  and  Resolving  Power  in  Television  Systems,  J.  Television  Soc., 
Vol.  2,  397-420  (June  1936). 

*  194.  RCA,  Radio  Facsimile,  Vol.  1  (October  1938). 

*  195.  V.  K.  Zworykin  and  G.  A.  Morton,  Television.    John  Wiley  <1940). 

*  196.  National  Television  Svstems  Committee,  Television  Standards  and  Practice  (1943). 

*  197.  C.  E.  Dean,  Television  Principles.    Hazeltine  Electronics  Corp.  (1944). 

Pulse  Modulation  for  Communication 

201.  R.  A.  Heising,  Transmission  System,  U.  S.  Pat.  1,655,543T  April  18,  1924-Jan.  10,  1928. 

202.  R.  H.  Ranger,  Reproducing  and  Transmitting  Pictures,  U.  S.  Pat.  1,848,839,  Feb.  26,  1924- 

March  8,  1932. 

'  203.  J.  L.  Finch,  Signaling  System,  U.  S.  Pat.  1,887,237,  May  31,  1929-Nov.  8,  1932. 
204.  R,  D.  Kell,  Signaling  System,  U.  S.  Pat.  2,061,734,  Sept,  29,  1934-Nov.  24,  1936. 
205. lW.  R.  Koch,  Secret  Communication  System,  U.  S.  Pat.  2,199,634,  June  21,  1938-May  7,  1940. 

206.  D.  G.  C.  Luck,  Signaling  System,  U.  S.  Pat.  2,227,596,  March  31,  1938-Jan.  7,  1941. 

207.  Bertram  Trevor,  Pulse  Signaling  System,  U.  S.  Pat.  2,361,437,  Dec.  4,  1940-Oct.  31,  1944. 

*  208.  E.  M.  Deloraine  and  E.  Labin,  Pulse-time  Modulation,  Electronics,  VoL  18,  No.  1,  100-104 

(January  1945). 
*209.  E.  M.  Deloraine  and  E.  Labin,  Pulse-time  Modulation,  Elec.  Com.,  Vol.  22,  91-98  (1944). 

210.  Editor,  Pulse  Position  Modulation  Technique,  Elec.  Ind.,  Vol.  4,  No.  12,  82-87,  180,  182,  184, 

186,  188,  190  (December  1945). 

211.  D.  D.  Grieg,  Multiplex  Broadcasting,  Elec.  Com.,  Vol.  23,  19-26  (March  1946). 

212.  F.  F.  Roberts  and  J.  C.  Simmons,  correspondence,  Wireless  Eng.,  Vol.  23,  93  (March  1946). 

*  213.  J.  J.  Kelleher,  Pulse-modulated  Radio  Relay  Equipment,  Electronics,  Vol.  19,  No.  5,  124-129 

(May  1946). 

*214.  D.  D.  Grieg  and  A.  M.  Levine,  Pulse-time-modulated  Multiplex  Radio  Relay  System-terminal 
Equipment,  Elec.  Com.,  Vol.  23,  159-178  (June  1946). 

215.  Editor,  Multi-channel  Pulse  Modulation,  Wireless  World,  Vol.  52,  187-192  (June  1946). 

216.  D.  I.  Lawson,  A.  V.  Lord,  and  S-  R.  Kharbanda,  Transmitting  Sound  on  the  Vision  Carrier 

of  a  Television  System,  J.  I.E.E.,  Part  III,  Vol.  93,  251-274  (July  1946). 

217.  W.  R.  Greer,  Pulse  Modulating  System,  Electronics,  VoL  19,  No.  9,  126-131  (September  1946). 

218.  D.  G.  Tucker,  Pulse  Distortion:  Interchannel  Interference  in  Multichannel  Systems,  J.  I.E.E., 

VoL  93,  323-334  (September  1946). 

219.  H.  S.  Black,  J.  W.  Beyer,  T.  J.  Grieser,  F.  A.  Polkinghorn,  A  Multichannel  Microwave  Radio 

Relay  System,  Elec.  Eng.,  VoL  65,  798-805  (December  1946). 

*  220.  RCA,  Television,  VoL  3  (December  1946);  VoL  4  (January  1947). 

221.  Airborne  Radar  Specifications,  Electronics,  Vol.  20,  132  (February  1947). 

222.  D.  G.  Fink,  Principles  of  Television  Engineering,  McGraw-Hill  (1947). 

*  223.  D.  G.  Fink,  Radar  Engineering,  McGraw-Hill  (1947). 

224.  Harvard  Univ.,  Electronic  Tubes  and  Circuits,  McGraw-Hill  (1947). 

225.  Reuben  Lee,  Electronic  Transformers  and  Circuits,  John  Wiley  (1947). 

226.  W.  M.  Goodall,  Telephony  by  Pulse  Code  Modulation,  Bell  Sys.  Tech.  J.,  Vol.  26,  395-409 

(July  1947). 

227.  R,  W.  Sears,  Electron  Beam  Deflection  Tube  for  Pulse  Code  Modulation,  Bell  Sys.  Tech.  J., 

VoL  27,  44-57  (January  1948). 

228.  L.  B.  Arguimbau,  Vacuum-tube  Circuits,  John  Wiley  (1948). 

229.  Australia,  Textbook  of  Radar,  Chapman  &  Hall  (1948). 

23D.  S.  Goldman,  Frequency  Analysis,  Modulation  and  Noise,  McGraw-Hill  (1948). 

231.  E.  C.  Pollard  and  J.  M.  Sturtevant,  Microwaves  and  Radar  Electronics,  John  Wiley  (1948). 

232.  R.  Q.  Smith,  Radio  Aids  to  Navigation,  Macmillan  (1948). 

233.  Denis  Taylor  and  C.  H.  Westcott,  Principles  of  Radar,  Macmillan  (1948). 

234.  C.  E.  Shannon,  A  Mathematical  Theory  of  Communication,  Bell  Sys.  Tech.  J.,  VoL  27,  379- 

423,  623-656  (July,  October  1948). 

235.  B.  M.  Oliver,  J.  R.  Pierce,  and  C.  E.  Shannon,  The  Philosophy  of  Pulse-code  Modulation, 

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239.  J,  H.  Dewitt,  Jr.,  and  E.  K.  Stodola,  Detection  of  Radio  Signals  Reflected  from  the  Moon 

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240.  W.  G.  Tuller,  Theoretical  Limitations  on  the  Rate  of  Transmission  of  Information    Proc 

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SECTION  10 
TRANSMISSION  CIRCUITS 


WIRE  TRANSMISSION  LINES 
ART  BY  JOHN  D.  TAYLOR  PAGE 

1.  Types  of  Communication  Transmission 

Circuits 02 

2.  Frequency  Spectrum 02 

3.  Electrical  Characteristics 02 

4.  Equivalent  Networks 08 

WAVE  GUIDES — THEORY 

BY   S.   A.   SCHELKUNOFF 

5.  Modes  of  Transmission 09 

6.  Propagation  Constants   of   Ideal  Wave 

Guides 10 

7.  Rectangular  Wave  Guides 11 

8.  Circular  Wave  Guides 13 

9.  Wave  Guides  of  Arbitrary  Cross-Section  15 

10.  Special  Characteristics  of  Wave  Guides  15 

11.  Wave-guide  Discontinuities 16 

WAVE-GUIDE  COMPONENTS 
BY  GEORGE  L.  RAGAN 

12.  Wave-guide  Characteristics 17 

13.  Flexible    Wave    Guides    and    Coupling 

Units 18 

14.  Wave-guide  Connectors 19 

15.  Bends,  Twists,  and  Angles 21 

16.  Impedance    Matching    and    Impedance 

Transformers 22 

17.  Transition  Units 24 

18.  Motional  Joints 26 

19.  Other  Components 28 


TRANSMISSION  IN  SPACE 

ART.  ^T  ^*   G.   SCHELLENG  PAGE 

20.  Wave  Propagation  and  General  View  of 

the  Radio  Spectrum 29 

21.  The  Ground  Wave 31 

22.  The  Sky  Wave 37 

23.  Obstacles  to  Transmission 42 

24.  Range  of  Radio  Stations  and  Broadcast 

Coverage 47 

MECHANICAL  FEATURES  OF 
TRANSMISSION  LINES 

BY  JOHN  D.  TAYLOR 

25.  Transmission-line  Construction 49 

26.  Electrical    Protection    of    Transmission 

Lines 58 

27.  Cable  Sheath  Corrosion 63 

COORDINATION  OF  COMMUNICATION 
AND  POWER  SYSTEMS 

BY  JOHN  D.  TAYLOR  AND 
HOWARD  L.  DAVIS,  JR. 

28.  Foreign  Wire  Relations 67 

29.  Structural  Coordination 68 

30.  Inductive  Coordination 73 

31.  Noise  Frequency  Induction 74 

32.  Noise  Induction  Mitigation 82 

33.  Low-frequency  Induction 88 


10-01 


TRANSMISSION  CIRCUITS 
WIRE  TRANSMISSION  LINES 

By  John  D.  Taylor 

1.  TYPES  OF  COMMUNICATION  TRANSMISSION  CIRCUITS 

Wire  communication  circuits  are  classified  in  two  groups,  open  (bare)  wire  and  cable. 
Each  group  includes  a  number  of  different  gages  and  types  of  wire,  and  the  electrical  char- 
acteristics of  each  gage  and  type  are  different.  Both  groups  have  their  respective  fields  of 
use,  and  frequently  these  fields  overlap  and  supplement  each  other.  Open  wire  is  generally 
economical  where  the  circuit  requirements  are  relatively  small  and  cable  costs  would  be 
prohibitive.  Cable  is  desirable  and  economical  where  large  groups  of  circuits  and  the 
higher-frequency  types  of  services  are  involved,  and  where  maximum  protection  from 
surface  interferences  is  essential. 

Cable  is  also  employed  for  specific  applications;  for  example,  coaxial  cable  for  television 
and  radio  transmission  lines. 

2.  FREQUENCY  SPECTRUM 

Communication  intelligence  and  signals  (with  the  exception  of  those  d-c  signals  used  in 
telegraph,  emergency,  control,  and  other  services)  are  usually  transmitted  between  points 
in  the  form  of  a-c  electrical  energy  or  wave  propagation  of  definite  frequencies  or  frequency 
ranges.  Because  of  the  different  devices  for  and  methods  of  generating  and  transmitting 
these  various  a-c  frequencies,  and  in  order  to  avoid  interference  between  the  numerous 
circuits  and  services,  frequency  allocations  have  been  determined  for  the  various  classes 
of  communication  and  signal  facilities. 

Communication  services  may  be  classified,  for  the  purpose  of  assigning  suitable  facilities, 
according  to  the  frequency  range  within  which  they  operate.  Figure  1  is  a  chart  of  fre- 
quencies employed  in  power  and  eornrminication  services  up  to  about  5  me.  (See  Sect.  17, 
art.  S.) 

3.  ELECTRICAL  CHARACTERISTICS 

Primary  constants  of  wire  lines  consist  of  resistance  r  in  ohms,  inductance  L  in  henries, 
capacitance  C  in  farads,  and  leakage  conductance  g  in  mhos.  For  convenience,  L,  C,  or  g 
may  be  expressed  in  smaller  units,  such  as  microfarads  (millionths  of  a  farad)  for  C. 
These  constants  vary  differently  for  different  wire  materials,  sizes,  spacings,  insulation, 
conductivity,  temperature,  and  moisture  conditions,  as  well  as  for  different  types  of 
construction,  as  indicated  in  Table  1. 

Secondary  constants  are  values  derived  from  primary  constants;  for  communication 
purposes,  they  consist  principally  of  (1)  impedance  z  in  ohms  and  phase  angle,  (2)  propaga- 
tion constant,  composed  of  the  attenuation  and  wave  length  constants,  (3)  wave  length, 
(4)  velocity  of  propagation,  and  (5)  cut-off  frequency.  From  the  various  constants,  the 
electrical  characteristics  of  the  different  wire  lines  have  been  determined,  and  these  char- 
acteristics are  useful  in  considering  the  suitability  of  such  lines  for  communication  purposes 
and  in  the  engineering  of  the  wire  communication  plant. 

The  secondary  constants,  expressed  in  terms  of  the  primary  constants,  are  as  indicated 
in  the  following  equations.  These  equations  apply  to  uniform  lines  of  infinite  length  or 
terminated  in  their  characteristic  impedances. 

CHARACTERISTIC  IMPEDANCE,  Z0  Z  0.  A  wire  transmission  line  has  both  series 
impedance,  z,  and  shunt  admittance,  y.  For  a  line  of  uniformly  distributed  primary  con- 
stants and  of  infinite  length  z  =  r  -f  JcaL  and  y  =  g  -f-  jtoC,  where  or  (angular  velocity)  = 
2x/ radians  [1  radian  =  (360/2*0  °}. 

ZQ  —  A/f  =  -\/ — '   ^^      ohms  (1) 

V*        V  +  oK?  ^; 

and 

r 
10-02 


ELECTRICAL  CHARACTERISTICS 


10-03 


&  « 

li 


IJ 


- 


•a  i 

a  ns 
<! 


+ 


-ill 


s 


«  o 


«j  m  o  —  CN  —  o  —  oo  --  u-\ooco-«J-<s 

r>^vr^OCT^r«^  t>iCO  —  o*«\t>.ooof>li>^ 

r^  vo  ^  o  -*•  ^  co  cc  >r  —  o  06  •«•  -^  o 


38SSSSS 

5  O  O  O  O  O  C3 


«Ar>»<<\"4-ooct4ci"\r*jr>.— 
—  O  <<v  ^S  **•  "^  cO  en,  -^  -<f 


OOOOOOO  OOOC2OOOOOO 


li 

— 


—  rsj  m  <N  in  PS  «^v 


3  S  O  O  O  O  O 
5  O  O  O  O  O  C3 


|||  §§|  g 


OOOOOOO       OOOOOOOOOO 


—  —  o^m  —  o 


eo  —  ^-  1^  cv  fs  t>,  oo 


2  fl<  C 

b  cb  - 
iA' 


:   :-  =1   i|   !|  = 


3      ° 

I     1 


<4-4       *-> 

0  'i. 

1  ! 


.a    -a 
-§ 


g44§f 


10-04 


TRANSMISSION  CIRCUITS 


For  standard  non-loaded  telephone  cable  lines,  L  and  g  are  relatively  small,  so  that  for 
approximate  computations  these  constants  may  be  neglected  and 

(2) 


—  /45°     ohms 


For  lumped  loaded  cable  lines  (the  usual  method  of  loading  such  lines),  the  midsection 
characteristic  impedance  ZQ  of  an  infinite  loaded  line  may  be  determined  from  a  pi  net- 
work, converted  to  a  T  network,  which  is  electrically  equivalent  to  the  combined  jumped 
and  distributed  constants  of  a  complete  loading  section  of  the  line.  The  a  (series)  and 


TELEPHONE 
SIGNALING 

STANDARD  FREQUENCY  FOR                FORMER 
RINGING  TELEPHONE   BELLS                STANDARD 
AND  SIGNALING  OVER  LOCAL             FREQUENCY  FOR 
TELEPHONE   UNES  AND  SHORT         RINGING  OVER 
TOLL  LINES    NOT  EQUIPPED               SHORT  TOLL 
WTTH  COMPOSITE    SETS                        LINES   t 

i                            * 

STANDARD  FREQUENCY   FOR   RING- 
ING   OVER    CARRIER  CIRCUITS  AND 
LONG  TOLL  LINES.  SIGNALING  CUR- 
RENT IS  INTERRUPTED  20  TIMES 

OPERATION  BY  VOICE  CURRENTS 

FREQUENCY 
IN   CYCLES 
PER  SECOND 

.  i 

L. 

(M       (U    1 

\          ^T       <f 

ool  1  1     8    1  , 

,1    '       '8 
8?    1 

1              !    1 
8§§       |     §   |§ 

*  T    T  r 

TELEPHONE 

f 

ORDINARY    TELEPHONE    TRANSMISSION 

PROGRAM    TRANSMISSION 

VOtCE-  FREQUENCY    TELEPHONE 

ABSOLUTE  LIMITS  OF  AUDIBILITY  EXTEND 
FROM  ABOUT  16  CYCLES  TO  32,000 
CYCLES,  BUT  ALL  ESSENTIAL  FREQUENCIES 
FOR  SPEECH  LIE  WITHIN  MUCH  NARROWER  BAND 

TELEGRAPH 

OEOtMARY  TELEGRAPH                                          CAREER    TELESRAPH 

AS  MANY  AS  to  TWO-WAY  VOICE-FREQUENCY 
NEUTRAL  AND  POLAR  SYSTEMS,.                       TELEGRAPH  CHANNELS  MAY  BE  CARRIED  ON 
GROUNDED  OR  METALLIC.  AL-                          BROAD-BAND  CARRIER  OR  FOUR-WIRE  CABLE 
THOUGH  THEORETICALLY  EMPLOY-                   FACIUTJES.  WHERE  BAND  WIDTH  OF  FACILITIES 
tNG  DIRECT  CURRENT,  TELEGRAPH                   IS  RESTRICTED,  FEWER  CHANNELS  ARE    POS- 
TRANSM*SSJON  INVOLVES  A-C                         SIBLE.  THESE    BLOCKS   MAY  BE  SUPERIMPOSED 
FREQUENCIES  RANGING  UPWARDS                   ONE  ON  TOP  OF  THE  OTHER   BY  DOUBLE   MOO- 
TO  ABOUT  25  CYCLES                                            ULATHDN  TO  UTILIZE  ANY  AVAILABLE  FREQUEN- 
(  *  1                                                       CY  RANGE 

f               1 

[ 

FREQUENCY 
IN  CYCLES 
PER  SECOND 

2         !! 

1 

3  S  f 

1    1 

:  §  s 

til             1 
SS88I      8    §  | 

1     ill             1 

§O     O    O  O  < 
1  §  III 

1  f  1 

i  1  1  o 

i  7* 

PO^^ 

?IAJJSff°  POW€R  TRAMS  -V^    STANDARD  FREQUENCY 
J£fSJ?!LWHE*'  ^TERNAT-              USED  MOST  EXTENSIVELY 
INC  CURRENT  RAILWAYS                    IN  POWER  WORK 

INC  IS  A  SECONDARY  CON~- 
Sl  DERATION 

FIG.  1.    Operating  Frequencies  for  Power  and 


ELECTRICAL  CHARACTERISTICS 

&  (shunt)  values  of  the  T  network  having  thus  been  determined, 


10-05 


(3) 


For  uniform  or  lumped  loaded  cable  lines  at  voice  frequencies  coL  is  large  with  respect  to 

r  and 


ohms 


(4) 


°1T<MJ1J?     f  777   TTTTTir     ?   ?7?  7 

{fti  !!!! 

**" 

n      "             LONG  -WAVE                    ^  f 
TRANSATLANTIC                  A 
TELEPHONE  QRCUITS          T 

fS^METERS-*s| 

BROADCASTING 

COAXIAL 

CABLE 

FIXED  AND  MOBI-LE    RADIO  TELEPHONE   BANt>S  AS  ALLOC 
AUTHORITY  THROUGH    THE  RANGE  FROM  tO  KILOCYCLES  T 

E    TO    W     AND     W    TO    E 

ATED  BY  GOVERNMENTAL 
3  30,000  MEGACYCLES 

RADIO   TELEPHONE                  L,. 

SHORT-WAVE  TRANSOCEANIC  '' 
TELEPHONE   CIRCUITS 
(4  TO  22  MEGACYCLES) 

>  CARR.IER   TELEPHONE 

TYPE  K  trninnnnnrgnmnn  CABL  \  SYSTEM 

1        2      3    4  5  6  78    10  12 
WTOE     ETOW              OPEN-WIRE    SYSTEMS 

TYPE  H 

ETO  WAND  WTOE 
TYPE  G            |       "'       |                                       W   TO   E         E    TO  W 

*  TYPE  jDOJODDDOODBO^ir 

WTOH     ETOW                                        1             6          12  12          1 

rrnf'rnr 

TVO^'   LJI                                                                           TYPE   M    |                                    | 

=  TOW            WTOE                                                             ^CA^IfS 

*TYPEC   |           111        11  fl       QDQ 
321213 

RADIO    TELEGRAPH 

F  XED  AND  MOBILE  RADIO  TELEGRAPH 
BANDS  AS  ALLOCATED  BY  GOVERNMENTAL 
AUTHORITY   THROUGH    THE  RANGE  FROM 
10  KILOCYCLES    TO  30,OOO  MEGACYCLES 

§                                    |    |||    | 

iiil  ||i} 

i 

NOTE:   LIGHTNING  FREQUENCIES  RANGE  FROM  A  FEW  HUNDRED  THOUSAND  TO  THE  HIGHEST 
FREQUENCIES  AND  ARE  INVOLVED  IN  THE  ENGINEERING  DESIGNS  OF  PROTECTION  IN 

BOTH  POWER  AND  COMMUNICATION   WORK 

*  THESE  REPRESENT   THE  FREQUENCY  ALLOCATIONS  FOR  ONE  SYSTEM  OF  THIS  TYPE  ; 
OTHERS  ARC   DISPLACED  SLIGHTLY, OR  EMPLOY  OTHER  SIDEBAND 

Communication  Services  (Courtesy  Bell  System) 


10-06  TRANSMISSION  CIRCUITS 

For  non-loaded  open  wire  lines  at  radio  frequencies 


2D 

ZQ  =  276  logic  —r     ohms  (5) 

a 

where  D  is  the  distance  between  the  centers  of  the  wires  and  d  is  the  diameter  of  the  wires- 
For  coaxial  (concentric  tube;  cable  lines  at  carrier  and  higher  frequencies 

Zg  ==  138  logio  -     ohms  (6) 

a 

where  a  is  the  outer  radius  of  the  inner  conductor  and  b  is  the  inner  radius  of  the  outer 
conductor. 

PROPAGATION  CONSTANT.    The  propagation  constant,  7,  is  a  function  of  the  series 
impedance,  2,  and  the  admittance,  y,  in  the  vector  relation 


-j«C)  (7) 

Also,  7  is  composed  of  an  attenuation  constant  a  and  a  wave  length  constant  /3  in  the 
relation 

7  =  a.  +  j8  =  y  cos  8  +  37  sin  9  (S) 

where  /  -  ,  - 

a  -  V  Va  vV  +  «W)tf  -f  or'C*)  +  1/2  («r  -  o>2LC)  (8a) 


0  »  V  Vs  "V  "(r2  +  ftWjtf  +  ^C2)  -  1/2  (&  -  <£LC}  (86) 

Equation  (S&)  gives  a  in  nepers  (1  neper  =  8.6S6  db)T  and  eq.  (86)  gives  £  in  radians 

(1  radian  —  57.296°).    It  is  usually  more  convenient  to  calculate  a  and  /?  from  eqs.  (7) 

and  (8)  than  from  eqs.  (8a)  and  (86). 

From  the  above  discussion  it  is  seen  that  the  propagation  constant  represents  both  the 

dying-away  and  phase-change  effects  of  the  voltages  and  currents,  as  they  progress  along 

the  line. 

If  y  is  given  for  1  mile,  then  the  total  attenuation  and  phase  change  effects  for  I  miles 

is  £7.    The  ratio  of  the  current  7s  at  the  receiving  end  to  the  current  Ji  at  the  sending  end 

of  a  uniform  line,  I  miles  long  and  properly  terminated,  is 

*j  =  8  -*y  =  e-i<«+#>  =  e-?«/^  (9) 

Thus,  the  magnitude  of  h  =  /i8~  la  and  the  two  currents  differ  in  phase  by  the  angle  Z£» 
which  is  expressed  in  radians. 
From  Eq.  (9) 

2.303  logiey  --  la 
h 

and  the  magnitude  of  the  propagated  current  I*,  in  terms  of  the  sending  current  Ji,  at 
any  point  along  a  uniform  transmission  line,  is 

h  „  l  (10) 


Ii        ,       -j     la 
10810    2J03 

Also,  if  P2  is  the  received  power  and  EI  and  h  are  the  sending  end  voltage  and  current, 
respectively,  with  phase  angle  B,  then 


t-*1*  cos  $  (11) 

showing  that  the  power  is  attenuated  in  accordance  with  the  square  of  the  current  ratio. 
If  the  constants  of  the  transmission  line  are  such  that  r/L  =  g/C  or  rC  —  Lg,  then  the 
line  attenuation  and  velocity  of  propagation  do  not  change  appreciably  with  frequency 
and  the  line  is  said  to  be  dwtortionfes»,  since 


For  standard  non4oaded  telephone  cables,  L  and  g  are  considered  negligible:  hence 
from  eqs.  (S)  _  ^^ 

a  ==  £  =  '\~-    nepers  (for  a}     and    radians  (for  £)  (13) 

For  uniformly  or  lumped  loaded  cable  lines  (up  to  about  mid-frequency  range)  where 
r  and  g  are  small  compared  to  uL  and  o><7  respectively,  an  approximate  expression  for 
o;  28 

£  +  i|       nepers  (14) 


ELECTRICAL   CHARACTERISTICS  10-07 

In  this  equation,  r/2L  and  g/2C  are  the  damping  constants  of  the  series  and  shunt  con- 
stants, respectively,  of  the  line. 

If  g  is  assumed  to  be  zero,  eq.  (14)  becomes 

_/"r 

nepers  (15) 

Equation  (15)  also  applies  for  open-wire  lines  in  the  r-f  range. 

Thus,  it  is  seen  from  eqs.  (13)  and  (15)  that,  for  values  of  (r/2L)  <  co,  a.  is  decreased  by 
increasing  L,  i.e.,  by  loading  the  cable  line,  although  the  full  advantage  is  not  realized, 
because,  in  loading  the  line,  r  is  increased  slightly  by  the  resistance  of  the  loading  coils. 

For  coaxial  (concentric  tube)  cable  lines  in  the  r-f  range 


(16) 

and,  where  both  conductors  are  of  the  same  material  and  the  line  dimensions  are  available, 

1/6)  .  1Q-,     neper  per  unit  length  (17) 


276  logio(6/a) 

where  p  =  resistivity  of  conductors  in  emu  (about  1730  emu  for  pure  copper). 
fi  —  magnetic  permeability  of  the  insulation. 
/  =  frequency  in  cycles  per  second. 
a  =  outer  radius  of  the  inner  conductor. 
b  —  inner  radius  of  the  outer  conductor. 

WAVE  LENGTH.  The  phase  of  the  voltage  and  current  for  a  uniform  line  is  contin- 
ually changing  in  a  modified  sine-wave  pattern,  as  progression  takes  place  along  the  line. 
A  complete  phase  change  of  2?r  radians  (360°)  will  occur  in  the  length  of  line  traversed  by 
the  voltage  and  current  during  the  time  they  pass  through  1  cycle.  Thus,  the  wavelength 
X  for  a  particular  line  may  be  equated  as 

2r 

X  =  -—     miles  (18) 

P 

&  being  the  phase  change  in  radians  per  mile. 

VELOCITY  OF  PROPAGATION.  Since  a  wave  length  is  the  length  traversed  by  the 
voltage  and  current  during  1  cycle,  the  velocity  of  propagation  W  may  be  equated  as 

W  =  X/    miles  per  second  (19) 

/  being  the  frequency  in  cycles  per  second. 

For  loaded  cable  lines,  the  inductance  and  capacitance  of  the  line  have  a  direct  bearing 
on  the  wave  length  constant,  wave  length,  and  velocity  of  propagation.  Since  r  and  g  are 
usually  negligible  for  such  lines,  eq.  (86)  for  /3  becomes 


|8  =  coVZc  =  2-n-fVLC  (20) 

Since  X  -  2w/fr 

97T/  1 

W  =  X/  =  —  -  =  —  =     miles  per  second  (21) 

0      VLC 

Note:  If  L  and  C  are  expressed  as  the  values  for  one  load  section,  then  /3  and  X  are  in 
loads  and  W  is  in  loads  per  second. 

CUTOFF  FRE  QUENCY.  For  non-loaded  lines  (cable  or  open  wire)  ,  the  cutoff  frequency 
is  usually  high  enough  for  all  practical  purposes  for  the  service  for  which  these  facilities 
will  be  used,  but  for  loaded  cable  lines  the  cutoff  frequency  is  an  important  factor  and  may 
be  expressed  as 

fc  -  —  ~  (22) 

irVLC 

where  L  is  the  inductance  of  the  loading  coil  in  henries  and  C  is  the  capacitance  of  the 
loading  section  in  farads  for  any  particular  line. 

Thus,  the  periodic  lumped  loaded  line  transmits  all  frequencies  up  to  a  critical  or  cutoff 
frequency.  However,  in  the  actual  line  there  are  some  deviations  from  the  ideal  loading 
and  some  resistance  in  the  line  and  loading  coils,  resulting  in  the  attenuation  of  the  trans- 
mitted frequency  range  to  some  degree. 


10-08 


TRANSMISSION  CIRCUITS 


4.  EQUIVALENT  NETWORKS 

Equivalent  networks  for  a  length  of  uniform  line,  transmitting  alternating  currents,  may 
be  constructed  in  the  T  or  pi  (TT)  form. 

For  the  T  network,  a,  the  value  of  each  of  the  two  series  arms,  and  6,  the  value  of  the 
shunt  arm,  may  be  obtained  from  the  equations 

(23o) 

&  =  -7^~r  (236) 

smh  ly 

where  ZQ,  I,  and  y  are  the  characteristic  impedance,  length  of  line  in  miles,  and  the  propaga- 
tion constant  per  mile. 


ly 
a  —  Z0  tanh  — - 


8 

--  10 
--8 

€ 
5 

--  5 

4 
3 

2 

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x* 

x 

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B.  £ 

r>d 

s 

1 

b2 

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-0.5 
-0.4 
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i  [ 

05          0.1            O.2     0.3 

0.5             1                23 

5 

10             20      30 

50 

Frequency  -  Megacycles  per  Second 

FIG.  2.    Calculated  Attenuation  vs.  Frequency  Characteristics,  600  ohm,  Two-wire  lines,  Solid  Copper 
Conductors ;  Leakage  and  Radiation  Neglected 

For  the  pi  network,  r,  the  value  of  each  of  the  two  shunt  arms,  and  s,  the  value  of  the 
series  arm,  may  be  obtained  from  the  equations 


r  =  jZ^o  coth  — 
2 

«  =  ZQ  sinh  Z-y 


(24a) 
(246) 


where  Z&  Z,  and  y  have  the  same  meaning  as  for  the  T  network. 

Figure  2  shows  the  calculated  attenuation-frequency  characteristics  of  600-ohm,  two- 
wire,  solid  copper  lines,  in  which  leakage  conductance  and  radiation  are  assumed  to  be 
negligible.  Calculations  are  based  on  the  equation 


=  S.686  - 


10  "*    decibel  per  mile 


(25) 


radii  of  the  various  conductors  in  miles,  /  =  megacycles  per  second,  and  ZQ  = 
ic  impedances  of  the  various  lines.    The  effect  on  current  distribution  of  the 
(important  only  when  the  ratio  of  separation  to  wire  diameter  is  less  than  20 
to  1)  is  neglected  in  the  equation. 


MODES  OF  TRANSMISSION 


10-09 


Figure  3  shows  the  calculated  attenuation-frequency  characteristics  of  copper  concentric 
tube  lines  having  optimum  ratios  of  conductor  radii.  Leakage  is  neglected.  Calculations 
are  based  on  eq.  (17).  Assuming  both  conductors  to  be  of  the  same  material,  a.  will  be 


1UU 

SO 

60 
50 

-f- 

J  

--  50 

40 
30 

20 

m     10 

-;  40 

JX 

x 

x* 

x^ 

<^^ 

x 

'* 

' 

x^ 

^ 

-;  10 

s     * 

1      6 

x 

1  

x"* 

--  8 

^x" 

x 

x 

.  _  & 

f*^"  — 

0.1 

25 

in 

•*' 

^ 

--  5 

1       4 

o      3 

J      2 

£ 

<       1 

r~n 

^i 

x 

^.x 

^ 

^ 

.X 

x 

.  ^ 

x 

L*" 

s<^ 

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X 

Ix" 

x< 

---0.25j 

in. 

x 

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x* 

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,,^ 

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X^ 

x^ 

x^ 

-" 

'x 

x-" 

x* 

x 

x'* 

x*^^ 

5  'in 

^ 

x* 

-I 

0.8 

0.6 
0.5 
0.4 
0.3 

0.-2 

0.1 
0. 

-_^X—  — 

~? 

"  — 

*-- 

-  -  Ct  ft 

x 

^ 

X 

x 

x 

x 

x^ 

•  -  0-Q 

x^1 

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<- 

-1 

5 

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x 

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s> 

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--03 

^ 

x' 

^x 

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-no 

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x'' 

X 

•'' 

.-  OJL 

05 

0.1            0.2     0.3        0.5 

1               23           5            '  10          ,   20    -.30 

50 

Frequency  -  -Megacycles  per  Second 

FIG.  3.     Calculated  Attenuation   vs.   Frequency   Characteristics,    Copper  Concentric  Tube   Lines. 
Optimum  ratio  of   conductor  radii.    Leakage  neglected.     Dimension  indicated  is  inner  radius  of 

outer  conductor. 


a  Tni-nimiiTn  when  the  ratio  b/a 
reduced  to 


3.6.    With  p  ~  1730  and  a  a  minimum,  eq.  (17)  can  be 


.  0.686V/ 


decibels  per  mile 


(26) 


where  /  is  in  megacycles  per  second  and  b  is  the  inner  radius  of  the  outer  conductor  in 
inches. 

WAVE  GUIDES— THEORY 

By  S.  A.  Schelkunoff 

Definition.  A  wave  guide  is  a  structure  consisting  of  either  conductors  or  dielectrics, 
or  both,  in  which  the  boundaries  between  different  media  are  cylindrical  (surfaces  made 
up  "by  a  translation  of  a  straight  line;  circular  cross-section  is  not  essential).  A  meted  wave 
guide  is  a  wave  guide  containing  at  least  one  conductor.  A  dielectric  wave  guide  is  a  wave 
guide  consisting  of  dielectrics  only. 


5.  MODES  OF  TRANSMISSION 

Three  types  of  waves  are  possible  in  straight  metal  wave  guides  filled  with  a  homogeneous 
dielectric:  transverse  electromagnetic  or  TEM  waves,  transverse  electric  or  TE  waves, 
and  transverse  magnetic  or  TM  waves.  In  TEM  waves  the  electric  and  magnetic  vectors 


10-10  TRANSMISSION  CIRCUITS 

are  perpendicular  to  the  direction  of  the  guide;  in,  TE  waves  the  electric  vector  is  so  dis- 
posed; and  in  TM  waves  the  magnetic  vector  is  perpendicular  to  the  guide.  TEM  waves 
are  possible  only  if  the  guide  consists  of  two  or  more  separate  conductors;  there  are  no 
restrictions  on  the  existence  of  the  other  types.  If  the  dielectric  is  non-homogeneous,  then 
in  general  the  waves  are  of  hybrid  type,  with  all  components  of  E  and  H  vectors  present. 

Each  wave  guide  permits  an  infinite  number  of  transmission  modes.  Besides  the  above- 
mentioned  general  characteristics,  each  mode  is  distinguished  by  a  transverse  field  pattern 
consistent  with  the  structure  of  the  guide.  Theoretically,  at  least,  it  is  possible  to  excite 
an  arbitrary  field  pattern  over  a  given  cross-section  of  the  guide;  but,  in  general,  this 
pattern  will  not  be  maintained  along  the  guide.  The  self-maintaining  patterns  are  the 
ones  that  define  the  various  modes  of  transmission.  In  general,  these  patterns  depend  on 
the  frequency  as  well  as  the  structure  of  the  wave  guide;  but,  if  the  dielectric  is  homoge- 
neous, then  the  patterns  are  independent  of  the  frequency  and  are  given  solely  by  the 
geometry  of  the  metal  boundaries.  Each  self-maintaining  pattern  is  either  attenuated  or 
is  traveling  with  a  phase  velocity  peculiar  to  it.  (See  Propagation  Constants,  article  6.) 
An  arbitrary  pattern  excited  over  a  given  cross-section  breaks  up  into  self-maintaining 
patterns.  The  field  at  various  distances  from  the  source  is  the  result  of  interference  be- 
tween the  self-maintaining  patterns  arriving  with  different  amplitudes  and  phases. 

In  wave  guides  with  rectangular,  circular,  and  elliptic  boundaries,  the  various  TE  and 
TM  modes  are  designated  by  a  double  subscript,  TEmn  and  TMmn,  where  m  and  n  are 
integers  appearing  in  the  mathematical  functions  describing  transverse  field  patterns. 
For  each  shape  of  the  guide,  indices  m  and  n  reflect  certain  physical  characteristics  of  the 
wave;  but  the  same  indices  for  different  shapes  correspond  to  waves  with  different  charac- 
teristics, and  waves  with  similar  characteristics  may  have  different  indices.  This  happens 
because  a  gradual  deformation  of  a  circular  boundary  into  a  rectangular  affects  different 
modes  differently. 

In  a  wave  guide  of  arbitrary  shape  the  various  modes  are  designated  by  a  single  subscript 
which  denotes  the  position  of  the  "cutoff  frequency"  on  the  frequency  scale.  The  dominant 
tease  is  the  wave  with  the  lowest  cutoff  frequency. 

6.  PROPAGATION  CONSTANTS  OF  IDEAL  WAVE  GUIDES 

For  an  ideal  non-dissipative  wave  guide  with  a  homogeneous  dielectric  the  propagation 
constant  yg  along  the  guide  for  a  given  transmission  mode  is 


yg  =  <ocVM£  \1  -  (  -  )  co  =  2*/  (1) 

\«C/ 

where  the  cutoff  frequency  fc  is  determined  by  the  permeability  ju  and  dielectric  constant  8 
and  by  the  geometry  of  the  boundaries.  For  /  <  /c,  y  is  real  and  the  wave  is  attenuated 
even  though  there  is  no  dissipation  of  energy.  For  /  >  fc  the  above  equation  becomes 


and  the  propagation  constant  is  imaginary.  The  wave  becomes  active  in  transmitting 
energy  to  large  distances  since  under  the  assumed  ideal  conditions  the  amplitude  of  the 
wave  does  not  diminish.  These  characteristics  are  the  characteristics  of  a  high-pass  filter. 

For  wave  guides  with  a  non-homogeneous  dielectric  there  is  no  simple  general  expression 
for  the  propagation  constant. 

PHASE  VELOCITY.    Above  the  cutoff  the  phase  velocity  along  the  guide  is 


'      Vl  -  <w«)*         =  V^  =  v£ 

where  v  is  the  intrinsic  velocity  of  the  dielectric  (velocity  of  light  in  the  dielectric),  c  is  the 
intrinsic  velocity  of  vacuum,  and  Zr  is  the  dielectric  constant  relative  to  vacuum.  In 
hollow  wave  guides  the  phase  velocity  is  always  higher  than  the  velocity  of  light  in  free 
space;  this  is  in  keeping  with  the  high-pass  characteristics  of  wave  guides. 

GROUP  VELOCITY.    The  group  velocity  or  the  velocity  of  a  "wave  packet"  is 


(3) 
Hie  product  of  the  group  and  phase  velocities  equals  the  square  of  the  intrinsic  velocity. 


RECTANGULAR  WAVE  GUIDES 


10-11 


WAVELENGTH  IN"  THE  GUIDE.    The  wavelength  \g  in  the  guide  (the  distance  from 
crest  to  crest  of  the  wave)  is 


(V/) 


Vl  - 


(4) 


where  Xr  is  the  free-space  wavelength  corresponding  to  the  given  frequency  /  and  ST  is  the 
dielectric  constant  relative  to  vacuum. 


7.  RECTANGULAR  WAVE  GUIDES 


FIELDS.  In  a  metal  wave  guide  the  effect  of  the  conductivity  of  the  walls  on  the  field 
distribution  is  negligible.  The  following  expressions  are  for  perfectly  conducting  walls, 
assuming  that  the  coordinate  system  is  disposed  as  in  Fig.  1.  The  time  factor  exp  jut  is 
omitted. 

TEmnt  wave  (if  traveling  in  the  positive  z  direction) : 


_          ,  nir        mitx   .    niry     ~ 

Ex  =  A  -r-  cos sin  — ; —  e    r 

b  a  b 

By  rr  E* 


KTE 


.  rrnr   .    WITTS         mry 

—  A  —  sin cos  —r—  e~'s 

o,  d  b 


A  XmrT         rmrx         mry 

A  - —  cos cos  — - —  e  " 

a  b 


J&ft 


(5) 


2  52 

where  m,  n  are  integers,  not  equal  to  zero  simultaneously.  The  quantities  g,  /z,  and  8  are 
respectively  the  conductivity,  the  permeability,  and  the  dielectric  constant  of  the  dielectric 
in  the  guide.  Figure  2  shows  electric  lines  for  some  TE  waves;  they  are  also  the  lines  of 
constant  Hz.  The  transverse  H  com- 
ponent is  perpendicular  to  the  E  vec- 
tor. The  density  of  lines  is  propor- 
tional to  the  transverse  field  compo- 
nents. The  pattern  of  the  TE1>0 
wave  in  Fig.  2  (a)  is  the  building 
block,  for  patterns  of  the  TEm,o 
waves  as  illustrated  in  Fig.  2(c); 


FIG.  1 


Electric  Lines  in  Rectangular  Guides;  (a) 

d) 


FIG.  2. 

TEi,o  Wave;  (&)  TEi.i  Wave;  (c)   TE2,0  Wave;  ( 
TEo,2  Wave 


m  is  the  number  of  such  blocks  in  the  pattern  of  the  TEw,o  wave.  In  the  TEo.n  wave  the 
lines  are  parallel  to  the  horizontal  faces  of  the  guide.  The  pattern  of  the  TEifl  wave  in 
Fig.  2(c)  is  the  building  block  for  the  pattern  of  the  TETO,R  wave  when  m  and  n  are  different 
from  zero ;  m  is  the  number  of  such  blocks  in  the  x  direction  (the  horizontal  direction) ,  and 
n  is  the  number  of  blocks  in  the  y  direction. 

TMw,n  wave  (if  traveling  in  the  positive  z  direction) : 


.  ttTT    .    WTTX        mry 

••  A  —  sin cos  — —  e" 

b  a  b 


mir         m-jrx    .     nicy 

—  —  A  —  cos sin 

a  a  o 


g  -f  .70)8  * 


(6) 


•      v 

where  m,  n  are  integers  not  equal  to  zero.  Figure  3  shows  magnetic  lines  for  some  TM 
waves;  they  are  also  the  lines  of  constant  E^.  The  transverse  E  component  is  perpendicular 
to  the  H  vector.  The  density  of  lines  is  proportional  to  the  transverse  field  components. 


10-12 


TRANSMISSION  CIRCUITS 


The  pattern  of  the  TMi.i  wave  in  Fig.  3  (a)  is  the  building  block  for  the  pattern  of  the 
TMm,n  wave;  m  is  the  number  of  such  blocks  in  the  x  direction,  and  n  is  the  corresponding 
number  in  the  y  direction. 

CUTOFF  FREQUENCIES.  The  cutoff  frequencies  for  the  various  modes  of  trans- 
mission are  the  frequencies  at  which  the  propagation  constant  vanishes;  at  these  fre- 
quencies the  propagation  constant  changes 
from  a  real  to  an  imaginary  value;  these 
frequencies  separate  the  pass  band  of  fre- 
quencies from  the  stop  band.  In  this  sense 
the  cutoff  frequencies  are  defined  only  for 
non-dissipative  wave  guides;  the  cutoff  fre- 
quencies of  slightly  dissipative  wave  guides 
are  usually  denned  by  neglecting  dissipa- 
tion. In  general,  the  cutoff  frequency  may 
be  defined  as  the  frequency  at  which  the 


Ita.  3. 


.  ta  ,  ^ctengular_Guides;  («> 


phase  of  the  propagation  constant  is  45°;  but  when  the  dissipation  is  really  large  the 
concept  of  "cutoff"  loses  its  practical  value. 

For  rectangular  wave  guides  the  cutoff  frequencies  and  corresponding  wavelengths  (in 
vacuum)  are 


V  U 


V(m/a)» 


where  c  is  the  velocity  of  light  in  vacuum. 
The  same  formulas  hold  of  TE  and  TM  waves.    For  the  dominant  wave 

Xi.o  =  2aVev        /i,o  = 


(7) 


(8) 


where  a  is  the  longer  dimension  of  the  cross-section  of  the  guide. 

ATTENUATION  IN  THE  PASS  BAND.  In  a  previous  section  the  exact  expression  is 
given  for  the  propagation  constant  y  when  the  walls  of  the  guide  are  perfect  conductors. 
To  allow  for  the  imperfect  conductivity  of  the  walls  the  following  correction  term  should 
be  added  to  y 

A-y  =  am(l  +  j)  (9) 

where  am  is  the  attenuation  in  nepers  per  meter  due  to  absorption  of  energy  by  the  walls. 
For  TEmn  wave 


0,  n 


(i+f' 

\  a       o 


0,  n  = 


«  0,  n 


(10) 


Vhere  p  —  b/a,  vmn  =  /»»„//,  and  ffi  is  the  surface  resistance  of  the  walls,  and 


(ID 


In  this  equation,  ^r  is  the  permeability  of  the  walls  relative  to  vacuum,  gm  is  the  conduc- 
tivity, and  \v  is  the  wavelength  in  vacuum  for  the  given  frequency  /.  For  pure  copper, 
g  —  5.80  X  107  mhos  per  meter  and 


(*  =  2.61  X  10- 
For  the  TMOT»  wave 


8.25X  H^  \ 


,. 

"  ^n  } 


(12) 


(13) 


!Hie  above  formulas  for  or,,,  break  down  in  the  immediate  vicinity  of  the  cutoff;  the 
attenuation,  does  not  go  to  infinity  —  it  is  merely  large  compared  to  the  attenuation  else- 
wfeere  in  the  pass  band  (see  a  later  section  on  the  attenuation  near  the  cutoff). 


CIRCULAR  WAVE  GUIDES 


10-13 


8.  CIRCULAR  WAVE  GUIDES 

FIELDS.    For  the  TEn,m  wave  the  cylindrical  components  of  the  field  vectors  are 


=  A  -  Jn 
p 


sin  n?  e 


=  A 


cos  r^  e 


where  kn,m  is  the  mth  non-vanishing  zero  of  the  first  derivative  of  the  Bessel  function 
Jnf(k),  and  a  is  the  inner  radius  of  the  guide.    Some  k's  are  given  below: 

&0,i  =  3.83  &o,2  =  7.02  &o,3  =  10.17 

&i.i  =  1.84  &i,2  =  5.33  Jfci.s  =    8.54 

&2,i  =  3.05  &2,2  =  6.71  &2,3  -    9.97 

h,i  =  4.20  kz.z  =  8.02  A3,3  =  11.35 

Figure  4  shows  electric  lines  for  some  TE  waves.    The  electric  lines  of  TEo.m  waves  are 
circles;  for  this  reason  TEo.m  waves  are  called  circular  electric  waves. 


FIG.  4.     Electric  Lines  in  Circular  Guides;   (a)  TEo.i  Wave;  (b)  TEi.i  Wave;  (c)  TE2,i  Wave;  (d) 
TEi,2  Wave;  (e)  TElfi  Wave  Between  Coaxial  Cylinders 


For  the  TMn,m  wave  the  field  is 

H0  =  A  -  Jn  (  -^-  ]  sin  n<p  e~ 


A  -  Jn 


cos  n<p  e 


(15) 


Ep  = 


=  -KZ™HP 


A/71775 


'    (g    +  /<* 


g 


<\W 


where  fcnm  is  the  mth  non- vanishing  zero  of  the  Bessel  function  /»(&).    Some  A's  are  givea 
below: 

fcotl  =  2.40        ^0,2  =  5.52         &o,3  =    8.65 

^L,I  =  3.83        ^1,2  =  7.02        fo.s  -  10.17 

fe,i  =  5.14         Ar2,2  =  8.42         fe,3  =  11.62 

Jfcs,i  =  6.38         &s,2  =  9.76         As,s  =  13.02 

Figure  5  shows  magnetic  lines  of  some  TM  waves.    The  magnetic  lines  of  TMo.m  waves 
are  circles;  for  this  reason  TM0,m  waves  are  called  circular  magnetic  waves. 


10-14 


TRANSMISSION  CIRCUITS 


FIG.  5.     Magnetic  lines  in  Circular  Guides;  (o)  TM0,i  Wave;   (&)  TMi.i  Wave;   (c)  TMit2  Wave; 

(d)  TM2,i  Wave 


CUTOFF  FREQUENCIES.    The  cutoff  frequencies  and  corresponding  wavelengths  (in 
vacuum)  are  _ 

(16) 


'  nm  J. 

n>nm 

where  c  is  the  velocity  light  in  vacuum  and  knm  is  the  quantity  denned  in  the  preceding 
section.    For  the  dominant  wave  ' 


l.7dVsr 


(17) 


where  d  is  the  inner  diameter  of  the  guide, 

ATTENUATION  IN  THE  PASS  BAND.  The  propagation  constant  7  given  in  a 
previous  section  is  for  the  case  of  ideal  non-absorbing  walls.  To  allow  for  absorption  by 
these  walls,  7  should  be  augmented  by 

A7  =  am(l  +  j)  (18) 

where  an  is  the  attenuation  in  nepers  per  meter  caused  by  the  absorbing  walls.    For  TE 
waves 


(18a) 


where  vnm  =  /n»//.     The  surface  resistance  (R  is  given  in  the  corresponding  section  on 
rectangular  wave  guides. 

For  circular  electric  waves  n  —  0  and 


,m2(l  -  *o,m2)-^  (186) 

If  the  dielectric  is  non-dissipative,  the  attenuation  of  circular  electric  waves  steadily 
decreases  with  increasing  frequency 
For  dominant  waves  ( 


[3.76(1  -  n,!2)-^  -  2.65(1  -  Ji.i2)^J10-3    neper/meter 


For  TM  waves 


'  (X  —    jrnjB* 


SPECIAL  CHARACTERISTICS  OF  WAVE  GUIDES      10-15 


9.  WAVE  GUIDES  OF  ARBITRARY  CROSS-SECTION 

CUTOFF    FREQUENCIES.     For  wave  guides  of  arbitrary  cross-section  the  cutoff 
wavelength,  (in  vacuum)  and  the  cutoff  frequency  are  given  by 


X.  = 


JJ 


'  JJlgrad  T|2  dS 


*-s 


(19) 


where  c  is  the  velocity  of  light  in  vacuum,  and  T  is  proportional  either  to  Ez  or  to  Hz* 
depending  on  the  type  of  the  wave.  This  formula  is  exact;  but  generally  it  is  not  feasible 
to  obtain  T  exactly.  However,  first-order  errors  in  T  lead  to  second-order  errors  in  \c, 
and  the  formula  is  extremely  useful,  especially  for  estimating  the  cutoff  frequency  of  the 
dominant  wave;  only  a  reasonable  guess  is  needed  for  T  (see  the  next  section). 

WAVE  GUIDES  WITH  CROSS-SECTIONS  OF  THE  TYPE  SHOWN  IN  FIG. 
6(a).  For  dominant  waves  in  wave  guides  of  the  shape  shown  in  Fig.  6 (a)  electric  lines 
run  across  in  the  manner  indicated  in  the  figure.  Such  wave  guides  are  essentially  pairs 
of  parallel  strips,  shunted  on  both  sides  with  cylinders,  Fig.  6(6).  Longitudinal  magnetic 


lines  run  largely  inside  the  cylinders,  in  opposite  directions.  On  account  of  the  increased 
capacitance  in  the  middle,  the  cutoff  frequency  of  such  a  wave  guide  is  lower  than  that 
of  a  rectangular  wave  guide.  To  make  an  estimate  of  the  frequency  we  let  T  —  Hz  —  +1 
in  one  cylinder  and  T  =  —  1  in  the  other.  Between  the  parallel  strips  we  assume  that  the 
longitudinal  magnetic  flux  varies  linearly  from  +1  to  —  1;  that  is,  we  let  T  —  1  —  (2iX/w), 
where  w  is  the  width  of  the  strips. 

The  gradient  of  T  is  zero  inside  the  cylinders  and  (— 2/tc)  between  the  strips.    Substi- 
tuting in  the  formula  contained  in  the  preceding  section,  we  obtain 


(20) 

where  h  is  the  distance  between  the  strips  and  5  is  the  cross-sectional  area  of  each  cylinder. 
In  the  above  form,  the  formula  applies  even  when  the  cylinders  are  not  circular.  The 
width  w  should  be  comparable  to  one-half  of  the  total  width. 

If  this  formula  is  applied  to  a  rectangular  wave  guide  whose  cross-section  is  broken  up 
as  in  Fig.  7,  we  find  Xi  ^  1.81  a V£T  instead  of  the  exact  value  Xi  =  SaVe*.  The  error  is 
about  10  per  cent.  The  error  diminishes  as  h  decreases  relatively  to  other  dimensions. 


10.  SPECIAL  CHARACTERISTICS  OF  WAVE  GUIDES 

CONDUCTION  CURRENT  IN  THE  WALLS  OF  A  WAVE  GUIDE.    The  conduction 

current  in  a  metal  wall  of  a  wave  guide  is  equal  to  the  tangential  magnetic  intensity  and 
is  perpendicular  to  it.  The  direction  is  that  in  which  a  right-handed  screw  will  advance 
when  the  H  vector  (which  is  supposed  to  be  attached  to  the  handle)  is  turned  through  90° 
to  make  it  coincident  with  the  normal  looking  into  the  wall. 

In  the  case  of  dominant  waves  in  a  rectangular  wave  guide  the  current  in  the  walls 
parallel  to  the  E  vector  is  strictly  transverse.  In  the  other  walls  the  longitudinal  current 
is  sinusoidally  distributed,  with  the  highest  density  in  the  middle;  the  transverse  current  is 
zero  in  the  middle  and  maximum  at  the  edges.  In  these  walls,  the  transverse  current  flows 
in  opposite  directions  from  the  middle.  As  the  frequency  increases,  this  current  diminishes. 


10-16  TRANSMISSION  CIRCUITS 

ATTENUATION  IN  THE  VICINITY  OF  THE  CUTOFF  FREQUENCY.  The  following 
expression  for  the  propagation  constant  is  valid  for  all  frequencies 

-y  =  W  +  fang  +  2a«-yo(l  +  f)  (21) 

where  70  is  the  propagation  constant  calculated  on  the  assumption  g  =  an  =  0.  Whereas 
am  is  infinite  at  the  cutoff,  amyo  is  finite.  For  frequencies  not  too  near  the  cutoff  this 
formula  is  unnecessarily  complicated,  and  it  is  recommended  only  for  the  immediate 
vicinity  of  the  cutoff.  _ 

CHARACTERISTIC  IMPEDANCES  AND  POWER  TRANSFER.  The  ratio 
Kt  =  Et/Ht  of  the  transverse  components  of  the  E  and  B  vectors  is  called  the  wave 
impedance.  The  average  power  W  transferred  per  unit  area  of  the  cross-section  of  the 
wave  guide  is 

W  -  K2Ht^  (22) 


where  Ht,en  is  the  effective  value  of  Ht  (the  factor  1/2  should  be  included  if  the  amplitude 
of  fft  is  being  used). 

For  the  dominant  wave  the  total  power  transfer  P  is  given  by 

P  -  Kp.vVet?  =  Kpjlen*  =  Kv.iVafctt  (23) 

where  Feff  is  the  effective  value  of  the  maximum  transverse  voltage,  7e«  is  the  effective 
longitudinal  current,  and  the  K's  are  the  characteristic  impedances.  For  rectangular 
wave  guides 


aVl  - 
For  circular  wave  guides 

764  ^  354 


. 

KP,i  -       ,  Ky,i  =      KP,vKp,z         (23a) 


(235) 


Vl  - 
For  wave  guides  of  the  shape  shown  in  Fig.  6,  approximate  formulas  are 

(23c) 


In  these  formulas  P  denotes  the  ratio  of  the  cutoff  frequency  for  the  dominant  wave  to  the 
operating  frequency. 

The  above  expressions  may  be  used  without  any  reservation  in  power  calculations.  The 
following  section  should  be  consulted  before  any  attempt  is  made  to  calculate  reflections 
and  standing  waves. 

11.  WAVE-GUIDE  DISCONTINUITIES 

Any  local  change  in  the  shape  of  a  wave  guide  or  any  obstruction  represents  a  wave-guide 
discontinuity.  Local  fields  are  likely  to  be  associated  with  such  discontinuities.  A  capacitor 
or  a  coil  shunted  across  a  low-frequency  transmission  line  are  examples  of  discontinuities. 
The  local  fields  store  energy  during  one  half  of  the  cycle  and  release  it  during  the  other; 
they  act  as  virtual  generators,  operating  on  power  borrowed  from  the  traveling  wave.  If 
the  operating  frequency  is  higher  than  the  cutoff  frequency  of  the  dominant  wave  but  lower 
than  the  cutoff  frequency  for  any  other  wave,  there  is  no  possibility  for  the  energy  to  be 
sent  back  to  its  source  or  to  the  load  in  any  mode  except  the  dominant.  In  these  circum- 
stances the  effect  of  the  local  field  made  up  of  higher-order  waves  may  be  represented  by  a 
reactive  transducer  which  in  its  turn  may  be  expressed  as  a  T  or  pi  network.  The  branch 
reactances  of  these  equivalent  networks  may  be  computed  in  terms  of  the  reflection  and 
transmission  coefficients  if  the  latter  are  measured  or  obtained  directly  from  a  solution  of 
an  appropriate  electromagnetic  boundary  value  problem.  In  some  special  cases  the  equiva- 
lent network  reduces  to  a  series  or  shunt  reactance. 

If  two  wave  guides  of  different  dimensions  are  joined  together,  the  junction  introduces 
a  reactive  discontinuity  of  the  above-mentioned  type  as  well  as  a  discontinuity  in  a  char- 
acteristic impedance  to  the  dominant  wave.  The  reflection  coefficient  will  be  determined 
not  merely  by  the  impedance  mismatch  but  by  the  reactive  discontinuity  as  well.  This 
is  a  general  statement,  and  it  applies  to  ordinary  low-frequency  transmission  lines,  for 
which,  however,  the  series  branches  of  the  reactive  discontinuity  are  so  small,  and  the 
sfemut  branches  so  large,  that  their  effect  is  negligible.  In  wave  guides,  on  the  other  hand, 
tike  e§€Ct  is  ordinarily  not  negligible. 


WAVE-GUIDE  CHABACTEEISTICS 


10-17 


Some  discontinuities  are  introduced  inadvertently,  as  in  joining  two  wave  guides;  others 
are  introduced  deliberately  as  circuit  elements.  Simplest  circuit  elements  are  transverse 
diaphragms  or  irises,  Fig.  8 (a),  (6),  (c),  and  transverse  strips  or  wires,  Fig.  B(d).  If  the 
irises  and  wires  are  thin,  they  are  substantially  shunt  reactances.  If  the  edges  of  the  iris 
are  perpendicular  to  the  E  vector,  as  in  Fig. 
8(a),  the  iris  is  capacitive;  if  the  edges  are 
parallel  to  the  E  vector,  as  in  Fig.  8(6),  the 
iris  is  inductive;  in  the  case  shown  in  Fig.  8(c) , 
the  iris  is  a  parallel  combination  of  an  induct- 
ance and  capacitance  and  may  be  designed 
to  be  an  antiresonant  circuit.  Thin  transverse  wires  introduce  a  shunt  inductance  nearly 
independent  of  the  frequency.  As  the  radius  increases,  the  series  branches  of  the  equivalent 
transducer  become  important. 


(a) 


(6) 


(d) 


FIG.  8 


WAVE-GUIDE  COMPONENTS 

By  George  L.  Ragan 


12.  WAVE-GUIDE  CHARACTERISTICS 

The  form  of  wave  guide  most  commonly  used  is  a  metal  tube  of  rectangular  cross-section 
wfyose  width  a  is  about  twice  its  height  b.  Within  a  twofold  frequency  range,  only  the 
dominant  (TEio)  mode  is  actively  propagated,  and  the  orientation  of  the  fields  within  the 
tube  is  unique.  That  is,  the  electric  field  is  perpendicular  to  the  broader  walls  of  the  wave 
guide.  By  contrast,  the  frequency  range  within  which  the  dominant  (TEn)  mode  in 
round  wave  guide  is  propagated  to  the  exclusion  of  all  others  is  only  1.31  to  lt  and  the 
orientation  of  the  fields  within  the  tube  is  not  uniquely  determined  by  the  wave-guide 
geometry  as  it  is  in  the  case  of  rectangular  wave  guide.  Consequently,  bends  and  irregu- 
larities in  round  tubing  cause  changes  in  field  orientation  and  even  introduce  elliptical 
polarization  effects.  It  is  because  of  this  difficulty  that  round  wave  guide  is  little  used  as 
a  transmission  line.  However,  round  wave  guide  is  frequently  used  in  short  sections  for 
rotary  joints.  In  this  application,  the  symmetrical  TMoi  mode  is  used. 

Expressions  for  calculating  cutoff  wavelength,  attenuation  in  copper  wave  guide,  and 
power  transmitted  in  common  wave-guide  modes  appear  in  Table  1. 

Table  1.     Wave-guide  Expressions  * 


Wave-guide  Shape 

Mode 

Xc 

a/ait 

P/Pit 

Rectangular            -  - 

TEio 

2a 

4.i6r-2-  +  ^Vi 

6  63a6 

Round 

TEn 

1.706d 

L2S  T  \\J  J 

•5     rc   f~n  jon  _i_    /    X\       | 

4.98d2 

Hound 

TMoi 

1.306d 

3.55^0.420+^  J 
2.72 

2.83*(^V 

\\j 

if  X  >  0.657d 

*  Tables  and  figures  in  articles  12-19  are  reproduced  by  permission  from  G.  L.  Ragan,  Microwave 
Transmission  Circuits,  Vol.  9,  Rad.  Lab.  Series,  McGraw-Hill  Book  Co.,  1948. 
f  a  =  attenuation  for  copper  wave  guide,  decibels  per  meter.  _4  2     /          ,^  ^z 

— 


The  notation  used  in  this  article  is  as  follows: 

a  =  larger  inside  dimension  of  rectangular  wave  guide,  in  meters. 

b  •==•  smaller  inside  dimension  of  rectangular  wave  guide,  in  meters. 

d  —  inside  diameter  of  round  wave  guide,  in  meters. 

X  =  free-space  wavelength,  in  meters. 
Xe  «  cutoff  wavelength  for  the  mode,  in  meters. 

a  =  attenuation  in  decibels  per  meter. 

P  =»  power  transmitted,  in  watts. 

E  =  electric  field  intensity,  in  volts  per  meter. 


10-18 


TRANSMISSION  CIRCUITS 


In  Table  2,  numerical  values  for  a  few  representative  wave  guides  are  given.  The  figure 
$max  =  30,000  volts/ cm  is  based  on  experimental  work  on  air  at  atmospheric  pressure 
(M.I.T.  Radiation  Laboratory  Series,  McGraw-Hill  Book  Co.,  Vol.  9,  Chapter  4). 

Table  2.     Characteristics  of  Representative  Wave  Guides 
(Courtesy  McGraw-Hill  Book  Co.) 


Army-Navy 
Designation 

Guide  Size 
CD,  in. 

Wall, 
in. 

Wave- 
length, 
cm 

Power,* 
mega- 
watts 

Loss.f 
db/m 
Copper 

Wavelength  J 
Band,  cm 

A.  Rectangular  (TEio  Mode) 

RG-48/U  
RG-51/U  
RG-52/U.     .. 

3        X  1.5 
1.25X  0.625 
1.0    X  0.5 
0.5    X  0.25 

0.080 
0.064 
0.050 
0.040 

10.  0 
3.2 
3.2 
1.25 

10.5 
1.77 
0.99 
0.223 

0.0199 
0.0725 
0.117 
0.346 

7.3  -13.0 
2.9  -  5.1 
2.3  -  4.1 
1.07-  1.9 

None  

B.  Round  (TEn  Mode) 

None 

3  ID 
I  OD 

10.0 
3.2 

16.6 
1.57 

0.0140 
0.0847 

10.0  -11.7 
3.18-  3.64 

None  

0.032 

*  Calculated  assuming  J^max  —  30,000  volts/cm. 
t  Calculated  values  for  copper. 

$  Based  on  maximum  wavelength.  10  per  cent  below  cutoff  wavelength,  minimum  wavelength  1  per 
cent  above  cutoff  wavelength  of  next  higher  mode. 

13.  FLEXIBLE  WAVE  GUIDES  AND  COUPLING  UNITS 

Flexible  wave  guide  is  used  similarly  to  flexible  coaxial  cable.     Applications  include 
connections  to  shock-mounted  units;  connections  to  pieces  of  equipment  which  must  be 


FIQ.  1.    Wound  Metal-hose  Wave  Guide  (Cour- 
tesy McGraw-Hill  Book  Co.) 


FIG.  2.    Titeflex  Wave  Guide  (Courtesy  McGraw- 
Hill  Book  Co.) 


moved  about;  temporary  or  emergency  replacement  lines;  and  as  patch  connections, 

particularly  in  test  equipment. 

Two  types  of  construction  which  have  proved  to  be  especially  useful  are  illustrated  in 

Figs.  1  and  2.    Figure  1  shows  the  * 'metal-hose"  type  manufactured  by  the  American 

Metal  Hose  branch  of  the  American  Brass  Co.,  Water  bury,  Conn.    This  type  is  made  of 

fairly  heavy  metal  wound  in  the  interlocking  manner 
indicated.  Flexibility  is  afforded  by  sliding  of  the  in- 
terlocked contacting  surfaces.  The  "Titeflex"  type 
shown  in  Fig.  2  is  manufactured  by  Titeflex,  Inc., 
Newark,  N.  J.  Titeflex  is  made  of  thin  metal  wound 
as  shown  and  soft-soldered.  Flexibility  is  afforded 
by  flexure  of  the  thin  metal  convolutions.  It  has 
been  found  that  a  molded-on  rubber  jacket  affords 
needed  protection  to  both  types  and  in  addition  im- 
proves the  performance  of  the  metal-hose  type  by 
causing  lower  contact  resistance.  A  complete  rub- 
ber-covered section  is  shown  in  Fig.  3.  Such  wave 

FIG.  3.    Rubber-covered  Flexible  Wave-    S^63  ,DMy  **   beat   °n  radii   ^ual   to   about   20 
guide  Assembly  (Courtesy  McGraw-Hill   tHnes  the  respective  nominal  dimensions  of  the  wave 
Book  Co.)  guide. 


WATE-GTJIDE   CONNECTORS 


10-19 


Several  other  types  of  flexible  wave  guide  have  been  found  useful,  especially  in  short 
lengths,  as  flexible  coupling  units.  These  include:  (a)  Corrugated  wave  guide;  essentially 
a  thin-walled  metal  bellows  of  rectangular  cross-section  (American  Metal  Hose) .  The  depth 
of  convolutions  must  be  a  small  fraction  of  a  wavelength.  (6)  Spun  bellows  assembly;  a 
soft-soldered  assembly  of  thin-walled  units  each  of  which  includes  a  circular  bellows  section 
the  depth  of  whose  convolution  is  effectively  one-half  wavelength  (American  Metal  Hose) . 


Flexible  cow 


Metal  hose  armor 


FIG.  4.     Typical  Vertebral  Assembly;  (a)  is  a  Single  Choke  Disk;  (fc)  is  the  Flexible  Cover;  (c)  is 
the  Assembly  (Courtesy  McGraw-Hill  Book  Co.) 


(c)  Cook  bellows  assembly  (Cook  Electric  Co.,  Chicago,  m.) ;  a  somewhat  more  rugged 
and  broader-band  bellows  similar  in  principle  to  (b) . 

A  coupling  unit  providing  a  maximum  flexibility  in  longitudinal,  transverse,  and  angular 
displacements  is  the  so-called  vertebral  type  illustrated  in  Fig.  4.  This  unit  is  based  on 
the  choke-flange  or  capacity-type  wave-guide  connector  described  in  article  14  below. 
Power  leakage  from  the  open  space  between  adjacent  sections  is  minimised  by  the  action 
of  the  choke  grooves,  and  fairly  large  displacements  may  be  tolerated  without  causing 
serious  impedance  mismatch. 


14.  WAVE-GUIDE  CONNECTORS 

Two  sections  of  wave  guide  are  usually  joined  by  couplings  of  either  contact  type, 
Fig.  5,  or  choke-flange  type,  Fig.  6.  The  contact  type  has  some  advantages  and  provides 
excellent  results  if  certain  precautions  in  design  and  use  are  observed.  The  choke-flange 
type,  however,  is  found  to  be  more  reliable  for  general  use. 


10-20 


TKANSMISSION  CIRCUITS 


The  design  of  contact  couplings  must  be  such  that  good  contact  is  secured  at  all  points 
around  the  periphery  of  the  butt-joined  wave  guides.  Particular  care  must  be  exercised 
to  avoid  the  formation  of  cavities  by  permitting  the  contact  to  be  made  at  points  on  the 
bolting  flanges  before  the  wave-guide  ends  are  forced  into  good  contact.  The  contact 
surfaces  must  be  carefully  machined  and  must  be  kept  free  of  dirt, 
corrosion,  and  mechanical  deformations. 

When  these  precautions  are  observed,  a  junction  is  obtained  which 
is  practically  perfect  in  all  respects.    This  type  of  coupling  is  ex- 
ceedingly valuable  in  certain  laboratory  design  work  where  the 
required  care  in  use  can  be  taken.    Likelihood  of  deterioration  of  the 
quality  of  the  contacts  in  use  under  field  conditions,  however, 
presents  a  serious  obstacle  to  their  general  utility.     One  contact- 
FIG  5     Contact  Coup-  tyl>e  coupling  which  has  given  satisfactory  service  for  use  with  1  !/2 
ling  (Courtesy  Me-  in.  by  3  in.  wave  guide  is  that  designated  by  the  Army-Navy  Cable 
Graw-HiU  Book  Co.)      Coordinating  Committee  as  UG-65/U  and  UG-66/U. 

The  choke-flange  type  coupling  of  Fig.  6,  in  which  quality  of  contact  is  unimportant, 
affords  a  connector  admirably  suited  to  use  under  adverse  conditions.  In  addition,  such  a 
coupling  scheme  is  very  useful  in  rotary  joints.  Parts  (b)  and  (c)  of  Fig.  6  are  designed 
principally  for  such  use.  The  wave-guide  ends  are  slightly  separated,  and  currents  inter- 
rupted by  the  gap  A  excite  a  folded  section  of  line  which  surrounds  the  gap.  This  section 
of  line  is  terminated  in  a  short  circuit  (closed  end),  and  its  length  is  effectively  one-half 
wavelength.  Hence  it  presents  at  its  input  end  essentially  zero  impedance  to  the  flow 
of  currents  interrupted  by  the  gap  A.  The  contact  between  flanges  occurs  at  the  point  B, 
which  is  at  the  midpoint  of  the  half-wavelength  line,  and  hence  at  a  point  of  essentially 
zero  current.  It  is  for  this  reason  that  the  quality  of  the  contact  between  flanges  is  un- 
important. 

Naturally,  there  is  only  one  specific  wavelength  for  which  the  conditions  outlined  above 
exist,  and  at  this  wavelength  only  is  the  connector  perfectly  matched  (i.e.,  reflectionless) . 
In  the  design  of  such  connectors,  one  must  be  guided  by  the  principles  discussed  below  if 
a  low  degree  of  frequency  sensitivity  is  to  be  achieved.  Since  the  same  principles  are  in- 
volved in  the  choke-type 
coupling  scheme  com- 
monly used  in  most  rotary 
joints  (see  article  17  be- 
low), they  will  be  dis- 
cussed in  some  detail. 

In  considering  the  ac- 
tion of  the  choke-flange 
coupling,  it  is  convenient 
to  consider  separately  two 
sections,  each  effectively 
one-quarter  wavelength 
long  at  the  design  fre- 
quency: (a)  the  radial  sec- 
tion from  A  to  Bt  and  (6) 
the  circular  groove  of 
depth  d.  The  circular 
groove  is  terminated  in 
zero  impedance  and  pre- 
sents at  B,  an  effective 
quarter  wavelength  away, 
a  high  impedance  which, 
in  series  with  the  contact 
resistance  at  Br  terminates 
the  radial  section  of  line. 
This  high-impedance  ter- 
mination of  the  radial  line 
is  transformed  to  a  low 
impedance  at  A,  an  effec- 
tive quarter  wavelength  away.  The  design  problem  resolves  itself  into  that  of  providing, 
at  frequencies  in  the  neighborhood  of  the  design  frequency,  a  maximum  impedance  at  B 
and  a  minimum  impedance  at  A.  It  is  easily  argued  that  these  objectives  are  realized 
(a)  by  maximising  the  characteristic  impedance  of  the  groove  section  by  making  the 
groove  width  x  as  large  as  is  practical,  and  (6)  by  minimizing  the  characteristic  impedance 
of  t&e  radial  Section  by  making  the  gap  y  as  small  as  is  practical 


FIG.  6.  Choke-flange  Couplings;  (a)  Rectangular  Wave  Guide;  (6)  and 
(c)  Circular  Wave  Guide  (Courtesy  McGraw-Hill  Book  Co.) 


BENDS,  TWISTS,  AND  ANGLES 


10-21 


For  example,  let  us  compare  the  voltage  standing  wave  ratio  (VSWR}  introduced  to 
X  =  9  cm  by  two  chokes,  both  perfectly  matched  at  10.7  cm,  differing  only  in  regard  at 
the  values  of  x  and  y.  For  one  design,  x  =  0.150  in.,  y  =  0.104  in-.,  VSWR  =  1.13.  For 
the  other  design,  x  =  0.250  in.,  y  =  0.050  in.,  and  VSWR  =  1.03, 

Design  details  and  performance  figures  for  a  number  of  choke-flange  units  are  given 
in  Table  3. 

Table  3.     Choke-coupling  Design  Details 
(Courtesy  McGraw-Hill  Book  Co.) 


Army-Navy  Type 
Choke  Flange 

Guide 
Dimen- 
sions, in. 

Choke  Dimensions,  in. 

Design 
Wave- 
length, 
cm 

Band 

Width  for 
r  -  1.05, 
% 

H  - 

y      \      d 

a 

b 

Rectangular-wave-gtdde  Choke. 
Fig.  6  (a)  —  TEio  Mode 

UG-54/U-53/TJ           .... 

2.84 
2.84 
0.90 
1.125 
0.900 
0.420 
0.420 

1.34 
1.34 
0.40 
0.50 
0.40 
0.170 
0.170 

4.015 
3.75 
1.  183 
1.332 
1.155 
0.501 
0.589 

0.250 
0.250 
0.063 
0.063 
0.125 
0.029 
0.063 

0.050 
0.030 
0.031 
0.031 
O.OIO* 
0.008 
0.008 

1.120 
0.865 
0.347 
0.347 
0.355* 
0.137 
0.156 

10.7 
9.0 
3.20 
3.20 
3.30 
1.25 
1.25 

±15 
±15 
±6 

U-200/U-2  1  4/U  

UG-40/U-39/U    

UG-52/U-51/U  

>±6 
>±2 

>±4 

UG-  11  7  /U-  11  6/U 

None  

c 

Circular-wave-guide  Chokes.     Fig.  6  (6)-—  TMoi  Mode 

None                    

0.4675 
1.187 

0.713 
1.479 

0.050 
0.093 

0.015 
0.030 

0.153 
0.312 

1.25 
3.30 

>d=4 
>±6 

None  

*  Designed  for  0.115-in.  separation  between  choke  and  flange. 


Lmearv 


15.  BENDS,  TWISTS,  AND  ANGLES 

If  rectangular  wave  guide  is  bent  or  twisted  gradually  enough,  i.e.,  in  a  length  represent- 
ing several  wavelengths,  the  reflections  set  up  in  the  wave  guide  are  negligibly  small.  Al- 
ternatively, it  may  be  bent  or  twisted  rather  abruptly  without  introducing  serious  reflec- 
tions if  the  mean  length  of  bend  or  twist  is  a  multiple  of 
half  the  guide  wavelength.  Care  must  always  be  exer- 
cised to  avoid  the  introduction  of  bumps  or  ripples  in  the 
wave-guide  walls  or  of  excessive  distortions  of  the  cross- 
sectional  dimensions.  The  achievement  of  this  end  is 
materially  assisted  by  filling  the  wave  guide  with  a  low- 
melting-point  alloy  such  as  Woods  metal  or  Cerrobend 
before  working  it,  and  melting  it  out  afterwards. 

Figure  7  illustrates  two  types  of  bend,  designated  as  H- 
plane  and  ^-plane  bends,  depending  on  whether  the  bend- 
ing radius  vector  lies  in  the  plane  of  the  magnetic  field 
lines  (IT)  or  electric  field  lines  (JB).  Incidentally,  these 
bends  are  sometimes  conveniently  referred  to  as  "hard 
bends"  (ff)  or  "easy  bends"  (E)  for  obvious  mechanical 
reasons.  The  performance  data  for  some  bends  and 
twists  are  given  in  Table  4. 

It  is  sometimes  preferable,  in  order  to  achieve  minimum 
space  factor,  to  substitute  fabricated  corners  for  bends. 
Two  types  of  well-matched  corner  are  illustrated  in  Fig. 
8.  The  double-mitered  type  is  usually  to  be  preferred  (a) 
because  the  dimensional  tolerances  are  larger,  and  (6) 
because  this  type  can  carry  more  power,  since  it  has  less 
acute  corners  and  a  less  restricted  cross-section.  Both 
types  may  be  designed  for  any  desired  angle,  not  being 
restricted  to  the  90°  type  drawn. 

It  is  found  experimentally  that  the  mean  separation  L 
between  miters  in  the  double-mitered  J?-plane  corner  is  a 
quarter  guide  wavelength  for  best  match.  This  is  easily  understood  as  providing  a  cancel- 
lation of  mismatches  by  reason  of  quarter-wave  spacing  between  identical  reflections.  In 
double-mitered  #-plane  corners,  however,  the  spacing  deviates  slightly  from  the  expected 
quarter-wave  value.  This  is  presumably  because  of  the  disturbance  due  to  a  greater 


FIG.  7.     Wave-guide  Bends;  (a) 

H  Bend;  (b)  E  bend  (Courtesy 

McGraw-Hill  Book  Co.) 


10-22 


TRANSMISSION  CIRCUITS 


Table  4.     Performance  of  Wave-guide  Circular  Bends  and  Twists 
(Courtesy  McGraw-Hill  Book  Co.) 


Type 

Wave-guide 
Size  ED,  in. 

Inside 
Radius 
Rt  in. 

e 

Design 
Wave- 
length, cm 

Band  Width 
for  r  below 
1-05,  % 

E-plane  bend 

1.34    X  2.84 

6 

45° 

10 

>±20 

1.34    X  2.84 

6 

90° 

10 

>±20 

1.125  X  0.50 

2 

90° 

3.3 

±  9 

0.90    X  0.40 

0.50 

180° 

3.3 

>±  9 

0.90    X  0,40 

0.25 

90° 

3.3 

>±  9 

0.90    X  0.40 

3.00 

90° 

3.3 

>±  9 

0.420  X  0.170 

0.50 

90° 

1.25 

>±  4 

fl-plane  bend 

1.34    X  2.84 

6 

45° 

10 

±10 

1.34    X  2.84 

6 

90° 

10 

>±20 

1.125X  0.50 

2 

90° 

3.3 

>±  9 

0.90    X  0.40 

0.192 

90° 

3.35 

±  4 

0.90    X  0.40 

1.1875 

90° 

3.30 

±  9 

0.420  ±0.170 

0.50 

90° 

1.25 

>±  4 

Type 

Wave-guide 
Size  ID,  in. 

Length,  in. 

Design 
Wave- 
length, cm 

Band  Width 
for  r  below 
1.05,  % 

Twists 

0.900  X  0.400 

2 

3.4 

±6 

0.900  X  0.400 

3 

3.4 

±3.7 

1.125X  0.500 

4 

3.3 

>±9 

0.420X  0.170 

U/4 

1.25 

>±4 

0.420X  0.170 

21/2 

1.25 

>±4 

(a) 


7 


$> 


X 


FIG.  8.     Wave-guide  Corners;   (a)   Double-mitered  Type;  (&)   Single-miter,  Cutoff  Type  (Courtesy 

McGraw-Hill  Book  Co.) 

excitation  of  local  fields.  Figure  9  gives  the  spacing  found  experimentally  to  give  well- 
matched  double-mitered  JS'-plane  corners  of  90°  total  angle.  Her  Xo/Xc  is  the  ratio  of 
design  free-space  wavelength  to  the  cutoff  wavelength.  The  band  width  (for  VSWH 
below  1.06)  is  found  to  range  from  about  20  per  cent  (occurring  for  Xo/Xc  of  about  0.6) 
to  #4x)ui  S  per  cent  (for  X0/X*  of  about  0.85). 


1&   IMPEDANCE  MATCHING  AND   IMPEDANCE   TRANSFORMERS 

In  designing  a  specific  wave-guide  component,  certain  parameters  are  varied  in  an  at- 
tempt to  arrive  at  a  design  which  performs  its  specified  function  and  at  the  same  time 
introduces  into  ihe  wave  guide  the  minimum  impedance  discontinuity,  i.e.,  the  minimum 


IMPEDANCE   MATCHING,   IMPEDANCE  TRANSFORMERS     10-23 


of  reflection  of  the  incident  r-f  wave.    It  is  frequently  impossible  to  achieve  the  desired 

degree  of  freedom  from  reflected  waves  without  sacrificing  quality  of  performance  in  some 

other  respect.    When  such  an  occasion  arises,  it  is  necessary  to  compensate  for  this  im- 

0.31  r- 


030 


<0.29 


0.28 


027 


0.50 


0.60 


0.70 


0,80 


0.90 


FIF.  9    Design  Curve  for   Double-mitered  Bi- 

plane   Corners    (Courtesy   McGraw-Hill   Book 

Co.) 


FIG.  10.     Wave-guide  Impedance-match- 
ing Diaphragms;  (a)  Inductive  Types;  (5) 
Capacitive  Types  (Courtesy  McGraw-Hill 
Book  Co.) 


pedance  discontinuity  by  introducing  into  the  wave  guide  an  impedance-matching  trans- 
former. 

Impedance  transformers  may  be  classified  into  two  general  categories,  fixed  and  variable. 

The  fixed  type  is  inserted  at  the  time  of 

fabrication  according  to  instructions 

given  by  the  designer,  whereas  the  vari- 
able type  may  be  altered  by  the  user  of 

the  equipment  to  achieve  the  desired 

performance.      Though    the    variable 

type  may  give  superior  performance 

when  adjusted  with  the  required  skill    * 

and  care,  it  is  capable  of  doing  more  K 

harm  than  good  if  improperly  adjusted. 

The  variable  type  is,  therefore,  to  be 

avoided  whenever  possible. 

The   fixed-type    impedance    trans- 
former usually  consists  of  one  or  more 

thin  metal  strips  soldered  into  the  wave 

guide  in  one  of  the  forms  shown  in  Fig. 

10.    The  most  widely  used  is  the  sym-    FlG*  • 

metrical  inductive  diaphragm  shown 

in  the  upper  right  figure.    The  equivalent  circuit  and  design  data  for  this  type  are  given  in 

Fig.  1 1.  The  asymmetrical  inductive 
diaphragm,  design  data  for  which  are 
given  in  Fig.  12,  is  used  somewhat 
less,  as  the  local  field  distortions  set 
up  by  it  are  more  extensive.  The 
capacitive  types  are  little  used  be- 
cause of  the  restriction  of  the  per- 
missible power  level  imposed  by  the 
high  electrical  fields  occurring  across 
the  gap  formed. 

The  curves  of  Figs.  11  and  12  are 
for  infinitely  thin  metal  strips.  A 
rough  compensation  for  increased 
thickness  may  be  made  by  assuming 
a  strip  to  extend  into  the  open  space 
by  an  amount  equal  to  half  the  thick- 
ness of  the  strip. 

FIG.  12.    Design  Curves  for  Asymmetrical  Inductive  Dia-        The  normalized  acceptance  B/Y« 
phragms  (Courtesy  McGraw-Hill  Book  Co.)  required  to  correct  for  a  measured 


10-24  TRANSMISSION  CIRCUITS 

voltage-standing-wave  ratio  r  may  be  determined  by  the  relation 

.   B        r-  I 


(1) 


Fo         Vr 

The  distance  d,  from  the  determined  position  of  a  minimum  in  the  voltage-standing-wave 
pattern,  toward  the  load  (for  inductive  diaphragms)  or  toward  the  source  (for  capacitive 
diaphragms)  may  be  determined  from  the  relation 

d_  _  90°  -  tan-^fe  J3/70) 

X,  "  720°  ^  } 

Alternatively,  both  B/YG  and  d  may  be  read  from  a  transmission  line  chart  or  admittance 
diagram. 

Several  other  types  of  fixed-impedance  transformer  have  seen  limited  use.  One  of  these 
is  the  quarter-wave  transformer  of  ordinary  transmission  lines.  A  wave-guide  section  of 
the  desired  lower  characteristic  impedance  is  formed  by  soldering  a  plate  of  suitable 
thickness  to  one  of  the  two  broad  walls  of  the  guide.  This  plate  is  the  full  wave-guide 
width  and  a  quarter  guide  wavelength  long.  Since  guide  width  a  is  not  changed,  charac- 
teristic impedance  is  proportional  to  unfilled  wave-guide  height  b. 

Another  scheme  utilizes  capacitive  "buttons"  soldered  in  place,  or  dents  formed  by  a 
suitable  rounded  tool.  Either  button  or  dent  forms  a  projection  from  the  broad  wall  of 
the  guide  which  acts  essentially  as  a  shunting  capacitive  element. 

Variable-impedance  transformers  or  "tuners"  have  appeared  in  numerous  forms.  They 
have  usually  either  two  or  three  adjustable  elements  appropriately  spaced  along  the  guide. 
Among  the  forms  which  the  adjustable  elements  have  assumed  are:  (a)  "Stubs,"  or  branch- 
ing sections  of  wave  guide  perpendicular  to  the  main  wave  guide  and  containing  adjustable 
short-circuiting  plungers.  The  stub  line  may  branch  from  either  the  broad  wall  (.EJ-plane 
stub)  or  narrow  wall  (H-pl&ne  stub)  of  the  guide.  (6)  Screws,  inserted  through  threaded 
holes  in  the  broad  wall,  (c)  "Slugs,"  obstacles  of  either  dielectric  or  metallic  materials, 
designed  to  alter  the  characteristic  impedance  of  the  guide  in  the  section  into  which  they 
are  inserted  (usually  a  quarter  guide  wavelength  long) .  These  slugs  are  ordinarily  inserted 
through  a  narrow  longitudinal  slot  in  one  of  the  broad  wave-guide  walls. 

Stubs  may  introduce  either  inductive  or  capacitive  susceptance.  Three  stubs  spaced 
at  quarter-wavelength  intervals  can,  in  theory,  transform  any  load  impedance  into  any 
desired  input  impedance.  In  practice,  if  one  is  interested  in  matching  out  or  introducing 
a  VSWR  of  2  or  less,  two  stubs  spaced  an  odd  number  of  eighth  wavelengths  apart  are 
adequate  and  have  the  advantage  of  easier  adjustment. 

Small-diameter  screws  inserted  into  the  wave  guide  introduce  a  capacitive  susceptance 
only.  This  limitation  makes  the  attainment  of  a  proper  adjustment  much  more  difficult 
than  that  for  stubs.  As  the  screw  diameter  is  increased  to  a  size  comparable  to  the  wave- 
guide dimensions,  it  achieves  an  inductive  effect  when  retracted,  just  as  does  a  stub,  which 
it  begins  to  resemble.  This  opens  up  the  possibility  of  a  screw-type  tuner  which  is  rela- 
tively easily  adjusted.  As  for  stub  tuners,  an  odd  number  of  eighth  wavelengths  proves  to 
be  a  good  spacing. 

A  single  screw  of  small  diameter,  mounted  on  a  sliding  sleeve  fitted  closely  around  the 
wave  guide,  and  projecting  through  a  longitudinal  slot  in  the  wave  guide,  constitutes  a 
very  useful  tuning  device.  By  adjusting  both  insertion  length  and  position  along  the  slot, 
any  tuning  requirement  is  easily  met. 

"Slug"  tuners  may  be  similar  in  action  to  the  single  screw  tuner  just  described.  Or 
they  may  consist  of  two  identical  slugs  whose  overall  reflection  is  varied  by  varying  the 
spacing  between  them,  and  the  phase  of  this  overall  reflection  is  varied  by  sliding  the  two 
slugs,  as  a  single  unit,  along  the  guide. 

A  simple  form  of  variable-impedance  transformer  is  the  phase  shifter  or  "line  stretcher" 
type.  This  type  does  not  alter  the  magnitude  of  the  wave  reflected  by  an  impedance 
discontinuity  but  merely  alters  its  phase.  Such  a  device  is  very  useful  in  promoting  sta- 
bility of  magnetron  oscillators  in  long  lines. 

One  common  form  is  made  simply  by  cutting  longitudinal  slots  in  both  broad  walls  of 
the  wave  guide  and  squeezing  the  section  thus  formed  to  alter  the  wave-guide  width.  As 
width  changes,  guide  wavelength  aad  hence  total  phase  length  of  the  section  change. 

17.  TRANSITION  UNITS 

Whenever  it  is  desired  to  couple  two  different  wave  guides  together,  either  in  the  same 
or  different  modes,  some  type  of  transition  unit  is  needed.  Similarly,  a  transition  between 
coaxial  lines  and  wave  guides  is  frequently  needed.  One  very  important  need  for  transition 


TRANSITION  UNITS 


10-25 


units  is  in  connection  with  rotary  joints,  where  the  dominant  wave-guide  mode  cannot  be 

used  (see  article  18). 

A  number  of  types  of  transitions  between  coaxial  lines  and  wave  guides  are  illustrated 

in  Fig.  13.    Types  (e)  and  (/)  are  especially  suited  for  low-power  work;  types  (g),  (d),  and 

(c)  are  recommended  for  intermediate  powers; 
and  types  (h)  and  (i)  are  especially  suitable 
for  high  powers. 

A  simple  transformer  designed  to  couple 
from  1  by  1/2  in.  wave  guide  to  1  1/4  by  5/8  in. 
wave  guide  is  shown  in  Fig.  14.  Such  a  trans- 
former section  may  be  calculated  by  making 


50J2 Coaxial  line 


X  (approx,) 


Probe  transitions 


Dimensions  used 

In  matching 
A-Iris  aperture 
D-End-plate  distance 
G-Couplfng  gap 
H-Contour  height 
.L-Loop  size 
P-Probe  insertion 
.R^lrfs  position 
iS-Coaxiaf  stub  length 


FIG.  14.    Quarter-wavelength  Transformer  be- 
tween 1  x  1/2  iftch  to  11/4  x  5/8  *&<&  Wave  Guides 
(Courtesy  McGraw-Hill  Book  Co.) 

the  intermediate  section  a  quarter  wavelength 
long  (in  terms  of  its  own  \g),  and  choosing  its 
dimensions  so  as  to  give  a  characteristic 
impedance  equal  to  the  geometric  mean  of 
those  of  the  joined  wave  guides.  For  this 
calculation,  the  characteristic  impedance  may 
be  taken  as 


8  X0  a 


(3) 


If  one  prefers,  two  rectangular  wave  guides 
could  be  joined  by  a  relatively  long  tapered 


FIG.  15.     Taper  from  Rectangular  to  Round  Wave 
Guide  (Courtesy  McGraw-Hill  Book  Co.) 

section.    Such  a  taper  is  commonly  used  be- 
tween rectangular  and  round  wave  guides,  as 
illustrated  in  Fig.  15.    If  space  does  not  per- 
mit such  a  taper  between  rectangular  and 
round  guides,  a  transformer  section  such  as 
that  shown  in  Fig.  16  may  be  used. 
Transitions  between  the  TEjo  mode  in  rectangular  wave  guide  and  the  TMoi  mode  in 
round  wave  guide  are  particularly  useful  in  rotary  joints.    Three  such  transition  units 
are  shown  in  Figs.  17,  18,  and  19.    Varying  degrees  of  complexity  are  illustrated.    In  Fig. 
17,  the  parameters  were  adjusted  so  that  satisfactory  impedance  match  and  TMoi  mode 


FIG.  13.    Transitions  from  Coaxial  Line  to  Wave 
Guide  (Courtesy  McGraw-Hill  Book  Co.) 


10-26 


TRANSMISSION  CIBCUITS 


purity  are  obtained  without  the  matching  diaphragms  or  mode-filter  ring  shown  in  the 
other  designs.  The  design  of  Fig.  IS  is  especially  recommended  for  high-power  work,  as 
no  sharp  corners  are  present  to  cause  breakdown. 


|xj*x  0.040* 

rectangular 

tubing  • 


FIG.  16.  Quarter  Wavelength 
Transformer  between  Round 
and  Rectangular  Wave  Guides 
(Courtesy  McGraw-Hill  Book 
Co.) 


+-  0.4675*10 
round  tubing 


0.158*-f 

FIG.  17.     Transition  from  Rectangu- 
lar TEio  to  Round  TMoi  Mode  for 
1.25  cm  Wavelength.  (Courtesy  Mc- 
Graw-Hill Book  Co.)    . 


Matching  windows 


FIG.     IS.     TMoi     Transition     for 
High-power    Use     (Courtesy    Mc- 
Graw-Hill Book  Co.) 


L.          ^-f 

Tx|'  Waveguide 

-^  in  2j 
t 

PoJyst 
M*ta 

xxv^a 

yrene  *• 
1  ring  " 

FIG.  19.     TMoi  Transition  with  Combina- 
tion Stub  and  Resonant  Ring  Filter  (Cour- 
tesy McGraw-Hill  Book  Co.) 


18.  MOTIONAL  JOINTS 

This  classification  includes  rotary  joints,  oscillating  joints,  hinge  or  "knuckle"  joints, 
and  universal  joints.  Rotary  joints,  exemplified  by  Figs.  20,  21,  and  22,  permit  continued 
rotation  about  an  axis;  oscillating  joints  oe^mit  limited  rotary  oscillations  about  an  axis. 


Secton  A-A 

Rotary  Joint,  Combining  Coaxial  Line  and  Wave  Guide,  for  Use  at  3-cm  Wavelength  (Cour- 
tesv  McGraw-Hill  "Rrw>v  rv  ^ 


MOTIONAL  JOINTS 


10-27 


Hinge  or  knuckle  joints,  Fig.  23,  permit  angular  displacements  of  one  wave  guide  with 
respect  to  another,  the  axis  of  the  hinging  being  perpendicular  to  the  wave-guide  axis. 
Universal  joints,  Fig.  24,  permit  the  motion  provided  by  gimbals. 


Type  N  coaxial  connector 


2.840  diam. 

Sern- 

cylinderical' 
end  plate 


2.56o"diarm 
/ 


;tchlng  Iris 


FIG.  22.     Rotary  Joint  Using 


IG.  22.        otary  Jont      sn 

TMoi  Mode  in  Round   Wave 

Guide  (Courtesy  McGraw-Hill 

Book  Co.) 


Flange 


Choke 


Waveguide 
1.340"x  2.840' 

inside 
dimensions 


Bearing 

Spring  "finger" 
contacts 


FIG.   21.      Rotary  Joint  for  10-cm  Wavelength 
(Courtesy  McGraw-HIU  Book  Co.) 


FIG.  23.    Hinge  or  Knuckle  Joints 
(Courtesy  McGraw-Hill  Book  Co.) 

Any  transition  unit  from  the  dominant  wave-guide  mode  to  a  coaxial  line  or  to  a  round 
wave  guide  excited  in  a  circularly  symmetrical  mode  may  serve  as  the  basis  for  a  rotary 
joint.  In  Figs.  20  and  21,  the  actual  rotation  is  accomplished  in  coaxial  line.  Gaps  in  the 
coaxial  line  conductors  are  bridged  by  "choke  couplings1'  similar  to  those  described  in 
article  14.  A  convenient  feature  of  the  high-power  rotary  joint  of  Fig.  21  is  the  inclusion, 
within  the  center  conductor  of  the  main  joint,  of  a  second  coaxial  rotary  joint  for  an 
auxiliary  transmission  line. 


10-28 


TRANSMISSION  CIRCUITS 


An  important  consideration  in  the  design  of  rotary  joints  using  the  TMoi  mode  is  con- 
cerned with  the  choice  of  a  length  L,  Fig.  22,  which  avoids  troublesome  resonance  effects. 
Such  resonances  are  associated  with  the  dominant  mode  fields  which  inevitably  exist, 
despite  efforts  to  avoid  their  excitation,  in  the  round  wave-guide  section. 
Any  rotary  joint  may,  obviously,  be  used  as  an  oscillating  joint  merely  by  restricting  its 

rotational  amplitude.  A  simpler  design,  however, 
may  be  arrived  at  by  applying  the  principles  of  the 
vertebral  assembly  of  chokes  and  flanges  described 
in  article  13. 

Two  constructions  of  #-plane  hinge  joints  are 
illustrated  in  Fig.  23.  Again,  the  choke  and  flange 
connector  is  used  as  a  basis  of  the  design.  Similar 
hinge  joints  of  $-plane  type  are  in  use. 

The  universal  type  joint  of  Fig.  24  is  essentially 
a  double  hinging  obtained  by  means  of  gimbals  and 
suitable  modifications  of  the  choke-flange  principle. 
In  this  design,  two  chokes  are  opposed,  rather  than 
choke  and  flange,  in  order  to  permit  greater  free- 
dom of  motion  without  leakage  of  power  from  the 
openings  around  the  joint.  The  antiresonance 
plugs,  shown  darkened,  prevent  trouble  from  reso- 
nances which  are  found  to  occur  whenever  two 
chokes  are  opposed.  The  same  resonance  trouble 
is  found  when  wave  guides  are  joined  by  opposing 
two  chokes. 


19.  OTHER  COMPONENTS 

The  foregoing  discussion  is,  of  course,  far  from 
exhaustive.  Space  has  permitted  the  inclusion  of 
only  a  few  designs  representative  of  the  types  dis- 
cussed. In  addition,  several  types  of  component 
have  not  been  treated,  even  cursorily,  because  of 
space  limitations. 

Among  the  more  important  omissions  are  the  so- 
called  T-R  or  duplexing  components  which  permit 
the  transmission  of  r-f  power  and  the  reception  of 
r-f  signals  on  the  same  wave-guide  and  antenna 
assembly.  Still  other  circuits,  which  are  con- 
spicuously omitted,  are  the  so-called  mixer  circuits 


0.500*x  1.000* x  0.050* 
wall  brass  tubing 


FIG.  24. 


Universal  Joint  (Courtesy  Mc- 
Graw-Hill Book  Co.) 


in  which  received  signals  are  mixed  in  suitable  crystals  or  tubes  with  a  local  oscillator 
signal  to  generate  the  i-f  signals  supplied  to  the  receiver. 

Those  interested  in  pursuing  these  omitted  items  will  find  them  described  in  the  appro- 
priate volume  of  the  M.I.T.  Radiation  Laboratory  Series  listed  in  the  bibliography. 


BIBLIOGRAPHY 

1.  Books  of  the  Massachusetts  Institute  of  Technology  Radiation  Laboratory  Series,  published  by 

McGraw-Hill  Book  CoM  New  York,  1947  and  1948. 
(a)  Vol.  9,  Microwave  Transmission  Circuits;  Ragan,  editor. 
(6)  VoL  17,  Components  Handbook;  Blackburn,  editor. 

(c)  Vol.  8,  Principles  of  Microwave  Circuits;  Montgomery,  Dicke,  and  Purcell,  editors. 

(d)  VoL  10,  Waveguide  Handbook;  Marcuvitz,  editor. 

(e)  Vol.  14,  Microwave  DupLexers;  Smullin,  editor. 
(/)  VoL  16,  Microwave  Mixers;  Pound,  editor. 

(#)  VoL  1,  Radar  Systems  Engineering;  Ridenour,  editor. 

2.  Microwave  Transmission  Design  Data,  Sperry  Gyroscope  Co.,  May  1944. 

3.  Reference  Data  for  Radio  Engineers,  Federal  Telephone  and  .Radio  Corp.,  Second  Edition,  1946. 

4.  MtcroiDawe  Techniques,  prepared  by  M.I.T.  .Radiation  Laboratory,  Bureau  of  Ships  Publication 


WAVE  PROPAGATION 


10-29 


TRANSMISSION  IN  SPACE 

By  J.  C.  Schelleng 


20.  WAVE  PROPAGATION  AND  GENERAL  VIEW  OF  THE 
RADIO  SPECTRUM 

Radio  propagation  is  a  special  instance  of  the  propagation  of  electromagnetic  waves. 
As  electromagnetic  waves  have  so  much  in  common  with  light,  an  understanding  of  radio 
propagation  begins  with  an  understanding  of  optics.  Radio  propagation  provides  ex- 
amples of  most  optical  phenomena:  interference,  reflection,  simple  refraction  and  double 
refraction,  diffraction,  etc.  Many  of  the  standard  formulas  of  optics  can  be  carried  over 
without  change  into  the  radio  field.  Although  it  is  beyond  the  scope  of  this  article  to 
discuss  these  fundamental  considerations,  a  few  general  principles  will  be  mentioned. 

RADIATION  AND  TRANSMISSION  IN  FREE  SPACE.  Radiation  of  electromagnetic 
waves  take  place  whenever  an  electrical  charge  is  accelerated.  The  wave  which  is  set  up 
is  transverse,  its  electrical  field  at  any  point  being  a  vector  perpendicular  to  the  direction 
of  transmission,  lying  in  the  plane  specified  by 
the  direction  of  propagation  and  the  direction 
of  acceleration  of  charge,  and  measured  in 
units  of  potential  per  unit  of  length  in  the 
direction  of  the  field,  e.g.,  microvolts  per 
meter.  When  the  space  surrounding  the 
source  is  free  of  material,  the  electrical  field  is 
propagated  outward  with  the  velocity  of 
light,  and  at  sufficiently  great  distances  it  is 
proportional  to  that  component  of  the  accel- 
eration which  is  parallel  to  the  electric  field  at 
the  point  in  space  under  consideration  (Fig. 
1).  These  statements  are  true  whether  the 
acceleration  is  sinusoidal  or  not.  In  radio 
communication,  the  accelerated  charges  are 
the  electrons  in  the  conductors  of  the  an- 
tenna. For  engineering  purposes  it  is  more  convenient  to  use  formulas  involving  quan- 
tities other  than  acceleration  and  charge.  Thus,  in  free  space,  the  electric  field  intensity, 
E,  from  a  short  electric  dipole  is 


Elect.  Fields 
Magn.  Field  - 


DIr.  of  Current 

and  Acceleration 

of  Charge 

FIG.  1.     Vector  Relations  in  Radiation 


8  =  60x  —  cos  6 
\d 


(1) 


ht  d,  and  X  are  the  effective  height,  distance  from  doublet  to  measuring  point  and  wave- 
length, all  in  the  same  unit  of  length  (e.g.,  meters);  8  is  in  volts  per  unit  length  (e.g.,  per 
meter),  and  7  is  in  amperes;  see  article  28,  pp.  5-52.  A  convenient  form  for  either  electric 
or  magnetic  dipole  is  _  .  * 


-  cos  9 


(2) 


in  which  P  is  the  radiated  power  in  watts,  and  8  and  d  are  in  the  same  units  as  for  eq.  (1). 
The  field  varies  inversely  with  the  distance.  For  an  especially  useful  formula  for  the  ratio 
of  power  picked  up  by  a  receiving  antenna  to  that  radiated  from  a  distant  transmitter, 
the  conditions  being  those  of  free-space  transmission,  the  reader  is  referred  to  Section  6, 
article  28,  eq.  (11). 

GROUND  "WAVE  AND  SKY  WAVE.  Two  general  modes  of  wave  propagation  are 
useful  in  radio  communication:  the  ground  wave,  which  passes  along  the  surface  of  the 
earth;  and  the  sky  wave,  which,  traveling  at  an  angle  with  the  surface,  passes  through  the 
lower  atmosphere,  is  reflected  from  the  upper  atmosphere,  and  is  enabled  in  this  way  to 
return  to  the  earth  at  a  distant  point.  The  ground  wave  is  used  over  short  distances;  the 
sky  wave,  or  ionospheric  wave,  is  required  for  the  longer  spans.  Intermediate  ranges  may 
involve  either  or  both,  depending  on  the  frequency  used,  the  time  of  day,  and  other 
circumstances. 

The  reciprocity  theorem  of  Lord  Rayleigh,  originally  derived  and  widely  used  for 
electric-circuit  analysis,  has  been  shown  by  Carson  and  others  to  be  true  in  cases  involving 
radiation  (B.S.T.J.,  April  1930,  and  earlier  papers).  It  results  from  one  form  of  this 
theorem  that  with  certain  limitations  the  efficiency  of  radio  transmission  in  opposite  direc- 
tions is  the  same,  provided  that  the  usual  measures  have  been  taken  to  match  the  generator 


10-30 


TRANSMISSION  CIRCUITS 


and  load  to  the  antennas,  and  that  the  path  is  free  of  elements  which  fail  to  act  reciprocally 
by  themselves.  A  one-way  amplifier  is  an  obvious  example  of  such  an  element.  Another 
example,  less  obvious,  actually  occurs  in  the  upper  atmosphere  itself,  namely,  the  ions 
which  tend  to  spiral  about  the  earth's  magnetic  field  in  one  direction  but  not  in  the  other, 
thus  producing  a  non-reciprocal  element.  As  a  result,  strict  reciprocity  can  be  expected 
where  the  ground  wave  is  concerned,  but  it  becomes  doubtful  with  the  sky  wave,  and  it 


almost  certainly  fails  where  rotation  of  the  plane  of  polarization  (indicating  a  magnetic 
effect)  is  observed.  Even  in  the  last  case,  the  averages  of  field  (as  opposed  to  instantaneous 
values)  usually  appear  to  be  reciprocal,  and  this  possibly  is  always  true  at  the  higher  fre- 
quencies. 

POLARIZATION.  For  frequencies  below  2000  kc  vertical  antennas  are  almost  uni- 
versally used.  This  is  primarily  because  it  is  usually  desirable  to  radiate  with  maximum 
efficiency  in  a  nearly  horizontal  direction,  which  is  relatively  easy  with  vertical  antennas 
"but  is  impossible  with  horizontal  antennas  unless  they  are  several  wavelengths  above  the 
ground.  Hence,  to  keep  antenna  dimensions  within  reasonable  limits,  horizontal  electric 
fields  are  not  used  except  for  short  waves. 

GEHERAL  VIEW  OF  THE  SPECTRUM.  With  respect  to  frequency,  daylight  propa- 
gation falls  into  natural  divisions.  These  may  be  listed  as  follows:  (1)  low  frequency,  long 


THE   GROUND  WAVE  10-31 

distances;  (2)  intermediate  frequency,  short  and  intermediate  distances;  (3)  "high  frequency, 
all  distances;  (4)  ultra-ionospheric  frequency,  short  distances.  Here  we  arbitrarily  describe 
a  short  distance  as  one  from  zero  to  100  miles,  an  intermediate  distance  as  one  from  100  to 
1000  miles,  and  a  long  distance  as  one  greater  than  1000  miles.  Likewise,  we  usually  think 
of  a  low  frequency  as  one  less  than  perhaps  500  kc.  Physically,  the  characteristic  that 
distinguishes  a  low  frequency  from  a  high  frequency  (3000  to  30,000  kc)  is  the  low  resis- 
tivity of  the  reflecting  layer  for  the  low  frequencies.  In  fact,  the  ionosphere  resembles  a 
fair  metallic  reflector  for  waves  of  low  frequencies.  The  wave  of  high  frequency,  on  the 
other  hand,  sees  in  the  layers  of  the  upper  atmosphere  something  like  a  reflecting  plane  of 
dielectric;  the  type  of  reflection  which  is  most  common  is  similar  in  many  ways  to  the 
familiar  optical  phenomenon  of  total  internal  reflection.  The  intermediate  frequencies 
form  a  transition  range  (500  to  3000  kc},  which  includes  much  ground-wave  transmission 
over  short  and  intermediate  distances.  Frequencies  commonly  designated  as  " ultra-high," 
but  which  are  better  called  "ultra-ionospheric"  because  they  are  not  reflected  by  the 
ionosphere,  are  those  greater  than  about  30,000  kc.  At  night  the  differences  between  iono- 
spheric waves  of  different  frequencies  are  much  less  marked  than  by  day.  Waves  having 
frequencies  in  excess  of  1000  megacycles,  more  or  less,  are  frequently  called  microwaves. 
See  also  Section  1,  article  24. 

Figure  2  gives  a  typical  overall  view  of  the  whole  radio  spectrum  for  distances  from  100 
to  10,000  miles  and  for  vertical  antennas.  Lines  are  drawn  indicating  the  distance  at 
which  a  radiated  power  of  1  kw  would  produce  certain  specified  field  strengths,  e.g., 
1  juv  per  meter.  Diagram  a  represents  transmission  conditions  on  a  summer  day;  diagram 
6,  those  on  a  winter  night.  On  winter  days,  the  sky  wave  becomes  generally  stronger 
than  on  summer  days.  As  a  general  rule,  for  a  given  distance  the  highest  frequency  that 
can  be  received  by  day  (the  skip  frequency)  is  greater  in  winter  than  in  summer;  and  the 
lowest  (absorption  limit)  is  lower  in  winter  than  in  summer.  At  night  lower  frequencies 
are  required  in  winter  than  in  summer.  Transmission  in  the  high-frequency  range  is 
markedly  affected  by  the  changes  accompanying  the  cycle  of  solar  activity  (e.g.,  sunspots) ; 
since  the  actual  phenomena  are  too  variable  to  be  represented  by  so  simple  a  chart  as 
Fig.  2,  the  comprehensive  data  and  predictions  issued  by  the  National  Bureau  of  Standards 
(Central  Radio  Propagation  Laboratory)  may  be  consulted  to  advantage  in  cases  of  actual 
use. 

21.  THE  GROUND  WAVE 

FREE-SPACE  TRANSMISSION.  Historically  the  propagation  of  the  ground  wave 
has  been  studied  by  examining  idealized  situations.  The  simplest  of  these  is  the  field  set 
up  in  free  space  by  a  simple  doublet  antenna.  Equations  (1)  and  (2)  give  the  appropriate 
solution,  and  apply  accurately  provided  that  the  earth  is  known  to  be  without  effect  and 
the  air  to  be  a  uniform  and  lossless  dielectric.  These  assumptions  are  not  true  in  general. 

PROPAGATION  OVER  PERFECTLY  CONDUCTING  PLANE  EARTH.  Some  situa- 
tions are  taken  care  of  by  assuming  the  earth  to  be  a  homogeneous  plane  and  then  applying 
the  standard  principles  of  optics.  This  is  particularly  simple  if  the  earth  in  effect  has 
infinite  conductivity.  The  solution  is  then  merely  the  combination  of  a  direct  wave  with  a 
reflected  wave  virtually  coming  from  the  "image"  of  the  antenna  in  the  earth  plane.  With 
low  antennas  and  infinite  earth  conductivity  we  are  led  to  a  simple  and  important  relation 
pointed  out  at  an  early  date  by  M.  Abraham.  For  distances  short  enough  not  to  violate 
the  assumptions  as  to  the  effective  flatness  of  the  earth  and  negligible  attenuation,  a  short 
vertical  grounded  antenna  of  effective  height  h  produces  a  field  strength  80  is-  the  region 
about  it  equal  to 

80  =  1207T  ^  cos  0  (3) 

a\ 

units  as  in  (1).    The  formula  corresponding  to  (2)  is 

V9OP         _  ,,_ 

80  =  ; COS  8  (4) 

a 

The  doubling  of  the  numerical  factor  in  passing  from  (1)  to  (3)  merely  expresses  the  fact 
that  the  field  may  be  regarded  as  the  sum  of  one  field  received  directly  and  another  by 
reflection  from  the  image.  Note,  however,  that  (2)  and  (4)  being  expressed  in  terms  of 
power  instead  of  current  have  factors  in  the  ratio  of  1  to  -v/2.  Formulas  (1)  to  (4)  apply 
for  distances  greater  than  a  few  wavelengths.  When  the  distance  is  of  the  order  of  1 
wavelength  or  less,  the  term  due  to  acceleration  of  charge  is  supplemented  by  terms  due  to 
velocity  and  position  (proximity)  of  charge,  but  for  most  practical  purposes  these  may  be 
neglected  for  distances  greater  than  1  wavelength.  Except  in  the  immediate  vicinity  of 


10-32  TRANSMISSION  CIRCUITS 

the  antenna,  the  velocity  of  phase  propagation  is  2.998  -  1010  cm  per  sec,  the  velocity  of 
light  in  air. 

PROPAGATION  ALONG  A  PERFECTLY  CONDUCTING  SPHERICAL  EARTH.  A 
next  step  beyond  that  represented  by  (4)  is  the  propagation  of  a  field  from  a  vertical 
grounded  antenna,  located  at  the  surface  of  a  perfectly  conducting  earth  which  is  spherical 
rather  than  plane.  Watson's  solution  [Proc.  Roy.  Soc.  (A)  Vol.  95,  S3,  546  (1919)]  is  in 
the  form  of  a  series  in  which  all  but  one  term  may  be  neglected  at  the  greater  distances. 
This  term  is  as  follows: 

J4 
&  =  0.11368o  ~^  e-°-oo376d/xH  (5) 

where  CQ  is  the  inverse  distance  field  as  given  by  (3)  or  (4),  and  d  and  X  are  in  kilometers,  d 
being  measured  along  the  surface.  This  formula  holds  when  d/\™  >  160.  For  smaller 
distances,  (3)  or  (4)  applies.  In  (5)  ,  atmospheric  refraction  has  been  neglected. 

The  following  simple  empirical  formula  based  on  Watson's  results  and  on  calculations 
of  other  terms  in  his  series  carried  out  by  C.  R.  Burrows  can  be  used  for  distances  less  than 
about  5000  km:  .  . 

8  =  8o(l  +  2s)^e-'  (6) 

where  z  =  0.0035  d/\^  and  d  and  X  are  in  kilometers.  All  these  diffraction  calculations 
assume  that  the  density  of  the  lower  atmosphere  is  independent  of  height;  atmospheric 
refraction  is  neglected.  When  the  deviation  of  refractive  index  from  that  at  ground  level 
is  a  simple  linear  function  of  height,  refraction  has  the  same  effect  as  increasing  the  size 
of  the  earth  to  a  virtual  radius  which  for  average  conditions  is  about  4/s  the  actual  radius. 
With  refraction  we  have  z  —  0.0029<i/X^-  It  is  not  obvious  which  of  these  values  should 
be  used.  Although  the  refraction  effect  is  important  at  sea  level,  it  must  become  small  at 
heights  of  several  miles.  Perhaps  the  effect  can  be  neglected  at  low  frequencies.  At  higher 
frequencies  experiments  clearly  show  that  it  should  be  taken  into  account. 

TRANSMISSION  ALONG  AN  IMPERFECTLY  CONDUCTING  PLANE  EARTH.  At 
distances  shorter  than  those  which  give  appreciable  attenuation  due  to  the  shadow  of  the 
bulge  of  the  earth,  strong  attenuation  due  to  energy  dissipation  in  the  ground  is  found  in 
many  cases  of  importance  in  practice.  Since  the  shadow  effect  is  then  unimportant,  the 
solution  of  wave  propagation  over  a  plane  of  finite  conductivity  becomes  applicable.  The 
basic  solution  of  this  problem,  due  to  A.  Sommerfeld  [Ann.  der  Physik,  (4)  Vol.  28,  665 
(19O9)3,  has  been  extended  by  various  investigators.  Owing  to  finite  earth  conductivity 
the  wave  ceases  to  have  the  simple  form  contemplated  in  connection  with  eq.  (3)  for  infinite 
conductivity,  the  inverse-distance  attenuation  of  which  represents  a  spherically  expanding 
wave.  The  field  strength  at  the  surface  decreases  in  intensity  more  rapidly  than  the 
inverse  of  distance,  owing  to  the  absorption  of  energy  by  the  currents  set  up  in  the  im- 
perfectly conducting  earth.  For  the  lower  frequencies,  an  approximation  due  to  van  der 
Pol  {Jahrbuch  der  Drahtloson,  Vol.  37,  No.  4,  152  (1931)]  is  useful.  This  is: 

o  (2  +  0.3p) 

' 


where  €Q  is  the  inverse-distance  field  given  by  (3)  or  (4)  and 

T  -  10~18      d 
P  =  ~~6~  '  M 

tr  is  the  conductivity  in  emu  units  and  varies  from  1  to  4  X  10  ~u  for  sea  water  to  10  ~15  for 
very  broken  land;  d  and  X  are  expressed  in  kilometers;  p  is  the  "numerical  distance"  of 
Sommerfeld.  Note  that  according  to  (7)  the  field  varies  inversely  as  the  first  power  of 
distance  near  the  transmitter  and  inversely  as  the  square  of  the  distance  for  p  ^>  20.  Also 
note  that,  since  X  and  cr  enter  only  as  the  product  <rX2,  the  field  strength  remains  unaltered 
when  the  wavelength  is  decreased  by  a  factor,  provided  that  the  conductivity  is  simulta- 
neously increased  by  the  square  of  that  factor.  Equation  (7)  holds  only  so  long  as  the  di- 
electric currents  in  the  earth  remain  negligible  compared  with  the  conduction  currents. 
This  is  insured  if  the  frequency  in  kilocycles  is  very  low  compared  with  1.8  X  10  ~l*  tr/K,  K 
being  the  dielectric  constant  of  the  ground.  Numerically,  the  frequency  should  be  con- 
siderably lower,  for  sea  water,  than  106  kc;  for  land,  than  104  kc;  and  for  fresh  water,  than 
250  kc,  the  figure  depending  on  the  ground  constants,  which  differ  from  place  to  place  and 
to  some  extent  with  temperature. 

An  interesting  characteristic  of  waves  traveling  along  an  imperfect  conductor,  first  dis- 
cussed by  J.  Zenneck  [Ann.  der  Physik,  (4)  Vol.  23,  846  (1907)],  is  that  parallel  to  the  sur- 
face there  is  a  longitudinal  component  of  electric  field,  that  is,  one  extending  in  the  direction 


THE  GROUND  WAVE 


10-33 


of  propagation.  Its  amplitude  and  phase  depend  on  the  resistivity  and  dielectric  constant 
of  the  ground,  the  phenomenon  being  very  different  over  fresh  water,  sea  water,  moist 
ground,  and  dry  ground.  Its 
amplitude  is  zero  for  perfect 
conductivity  and  increases  as 
the  conductivity  decreases. 
The  phase  of  this  component 
in  general  differs  from  that  of 
the  vertical  component  so 
that  in  the  vertical  plane  the 
wave  exhibits  elliptical  polari- 
zation. By  Poynting's  the- 
orem the  existence  of  this 
component  is  a  necessary  ac- 
companiment of  energy  loss 
in  the  ground.  The  phenom- 
enon is  of  importance  in  the 
design  of  wave  antennas, 
which  depend  for  their  effec- 
tiveness entirely  on  this  hori- 
zontal component.  [See 
"Wireless  Waves  at  the 
Earth's  Surface"  by  G.  W.  O. 
Howe,  Wireless  Engineer,  VoL 
17,  385  (September  1940).] 

IMPERFECTLY  CON- 
DUCTING SPHERE.  A  still 
closer  approximation  to  ac- 
tual conditions  is  afforded  by 
the  assumptions  that  geomet- 
rically the  earth  is  a  perfect 
sphere  (i.e.,  that  local  irregu- 
larities may  be  ignored)  and  J 
that  electrically  over  any 
given  path  it  has  uniform  con- 
ductivity and  specific  induc- 
tive capacity.  As  to  refrac- 
tion in  the  atmosphere,  the 
assumption  made  above  in 
connection  with  a  perfectly 
conducting  sphere  is  repeated. 
Though  these  assumptions 
still  describe  a  somewhat  ide- 
alized picture,  they  neverthe- 
less represent  a  large  step  in 
the  direction  of  realism,  a  dif- 
ficult and  important  mathe- 
matical task  which  has  been 
successfully  accomplished  by 
the  combined  labors  of  several 
investigators,  including  Balth 
van  der  Pol.  Excellent  sum- 
maries, including  bibliogra- 
phies, have  been  given  by  C« 
R.  Burrows  and  M.  C.  Gray 
[Proc.  I.RJE.,  Vol.  29,  No.  1, 
16  (January  1941)]  and  by  K 
A.  Norton  [Proc.  I.R.E.,  Vol. 
29,  No.  12,  623  (December 
1941)].  Figures  3,  4,  and  5 
are  based  on  the  paper  of 
Burrows  and  Gray. 

Figure  3  gives  theoretical 
field  strengths  between  two 


100 
80 
60 
40 
20 
0 
-20 
-40 

100 
80 

0 

|    60 

> 

2-40 

1    20 

1 

0           0 

a 
-20 

^ 

^*s^ 

1 

I 

\ 

^ 

>i^> 

^x"" 

xX 

Freq 
me 

uen 

lac* 

cy  In 
cles 

1 

Vertlca    | 
polarization 
Poor  soil 

x 

X 

x 

*s 

>>Hx 

^ 

x£ 

X 

x 

s 

-IX 

X 
X 

x 

>J_V_ 

X 

"X 

\ 

X 
x 

N 

s      f            ^S 

k     N 

\ 

X 

x 

K 

N:^\ 

N 

\ 
\ 

\ 

\60         ' 

v!iX 

^\ 

\ 

^ 

s 

\ 

,600    \ 

\l      < 

\ 

\ 

\ 

\ 

\ 

""•*», 

^x^ 

^ 

\ 

"^ 

x 

•*4, 

N5^^fc^ 

Vertical 
polarization 
Good  soil 

x 

x 

x 

x. 

X 

X 

Sk; 

X 

x 

x 

X 

x 

s    s 

|bx 

x 

2 
\ 

x 

X 

Tx 

x 

x 

\ 

S 

"S 

0. 

\ 

6  \ 

^ 

I  "^ 

X 

\f 

\ 

\ 
\ 

\ 

\ 

\€ 

•  \ 

\ 

y 

\ 

60CJ 
\\ 

\ 

\ 

\1\ 

\  i 

\ 

100 
80 
60 
40 
20 
0 
20 

-40 
0 

""^x 

^^ 

^ 

^L"** 

--,. 

X 

Tt^. 

Vertical 
polar  zatlon 
Sea  water 

X 

s, 

•«, 

-<5: 

rk 

x 

X 

X 

\ 

^ 

^ 

X^-2 

( 

x 

N 

\ 

\ 

^ 

\: 

.5\^ 

\ 

f 

\S 

\ 

N600 

\ 

\ 

\ 

\ 

1 

\ 

512             5 

10       20          &0 
Distance  In  miles 

100    200        500  1000 

FIG.  3.  Field  Along  Imperfectly  Conducting  Spherical  Earth,  with 
4/3  Earth-radius.  Short  vertical  antenna  at  ground  level,  measure- 
ment at  ground  level,  for  sea-water  <r  =  4:  X  10"11  emu,  €  =  80; 
good  soil  or  =  2  X  10  ~13  emu,  «  =  30;  poor  soil  <r  =  10  ~14  emu, 
e  =  4,  1  kw  radiated  (Courtesy  Proc.  I.R.E.) 


points  on  the  surface  of  the  earth  when  vertical  polarization  is  used.    "Standard  refraction" 


10-34 


TRANSMISSION  CIRCUITS 


conditions  (4/3  earth  radius)  are  assumed  to  occur  over  a  sufficient  depth,  of  atmosphere  to 

produce  bending  at   all 

100  | — *=vT — 1    Mini ! ! mn 1      I    .^I.l.U'n       frequencies,  a  postulate 

frequently  made  which 
might,  however,  prop- 
erly be  questioned  with 
reference  to  the  lower 
frequencies  shown,  since 
with  them  transmission 
to  the  longer  distances 
involves  at  intermediate 
points  regions  which  are 
far  above  the  earth's 
surface  and  at  which  the 
rate  of  decrease  of  re- 
fractive index  with 
height  is  no  longer  com- 
parable with  that  at  the 
surface.  The  legend  of 
the  figure  gives  the  soil 
constants  assumed  in 
the  calculation.  Impor- 
tant features  of  the 
curve  are  the  "inverse- 
distance"  tendency  at 
the  upper  left  corners  as 
in  the  approximation  of 
eq.  (4),  the  rapid  drop 
due  to  failing  diffraction 
at  the  lower  right  as  sug- 
gested by  eq.  (5),  and 
the  inverse  square  of 
distance  in  between  as 
indicated  by  eq.  (7)  (for 


-20 


Distance  in  mifes 


100 


1000 


Height  sain  In  decibels 
H*  M  w 
o  o  o  o 

^ 

- 

^ 

^ 

^ 

^ 

15( 

\  M 

c 

—  - 

—  - 

,—  - 

,^-" 

>                  10                 20         30           50                100              20 
Antenna  heighi  to  feet 

FIG.  4.     Fields  at  150  Me  as  a  Function  of  Height  of  Transmitter  and 
Receiver,  Good  Soil  (Courtesy  Proc.  I.R.E.) 


100 


-20 


foorjsoH- 
5ea  Water 


10 


I  polarization 

'  -Poor  soil  I 

•Good  soil] 

•Sea  water 


p  }§>  20).  A  useful  series  of  calculations  along  these  general  lines  (K.  A.  Norton)  is  included 
in  "Standards  of  Good  Engineering  Practice  Concerning  Standard  Broadcast  Stations/' 
issued  by  the  Federal  Communi- 
cations Commission,  for  sale  by 
the  Superintendent  of  Docu- 
ments, Washington  25,  D.  C. 
This  document  also  gives  com- 
prehensive information  of  the 
ground  conductivities  pertinent 
in  this  frequency  range  in  the 
form  of  a  United  States  map. 

Figure  4  illustrates  propaga- 
tion from  one  point  on  or  near  the 
ground  to  another  at  heights  from 
zero  to  40,000  ft  above  it  for  the 
special  case  of  150  megacycles. 
If  the  transmitting  antenna  (ver- 
tical polarization)  is  on  the 
ground,  the  upper  graph  gives  the 
calculated  field  directly.  The 
lower  graph  gives  the  correction 
to  be  added  if  the  antenna  is  ele- 
vated not  more  than  200  ft.  The 
increase  which  accompanies  the 
elevation  of  the  receiver,  and  the 
inevitable  failure  of  diffraction  at 


50 

Distance  In  miles 


greater    distances,    are    striking 
features  of  the  graph. 

Figure  5  exemplifies  for  fixed 
antenna  heights  and  fixed  fre- 
<atteucy  how  the  field  depends  on  the  underlying  ground  and  on  the  polarization.    Note  the 
eness  of  horizontal  polarization  to  ground  constants. 


FIG.  5.    Fields  at  150  Me  and  10,000  Ft.  as  a  Function  of 
Ground  and  Polarization  (Courtesy  Proc.  I.R.E.) 


THE   GROUND  WAVE  10-35 

ULTRA-IONOSPHERIC  RANGE.  This  is  the  range  of  frequencies  higher  than  those 
capable  of  reflection  by  the  ionized  upper  atmosphere.  The  dividing  frequency  is  not  at 
all  sharp  or  constant  but  is  of  the  order  of  30  megacycles.  Ultra-ionospheric  waves  whose 
length  is  short  enough  for  the  practical  application  of  quasi-optical  techniques  such  as  the 
use  of  parabolic  reflectors  and  lenses  have  been  called  "microwaves,"  and  here  an  equally 
hazy  division  might  be  placed  somewhere  near  1000  megacycles. 

Although  the  absence  of  reflection  of  ultra-ionospheric  waves  might  hypothetically  be 
accounted  for  by  their  penetration  into  a  region  where  they  are  dissipated  by  absorption 
before  being  freed  by  reflection,  three  experiments  now  indicate  actual  passage  through 
the  ionosphere  into  interstellar  space.  In  chronological  order  these  are  the  detection  of 
galactic  noise  by  K.  Jansky  (and  the  subsequent  mapping  of  the  Milky  Way  by  Grote 
Reber),  the  measurement  of  thermal  radiation  from  the  sun  by  G.  C.  Southworth 
[J.  Franklin  Inst.,  Vol.  239,  No.  4,  285  (April  1945)],  and  the  dramatic  "detection  and 
ranging"  of  the  moon  by  radar  reported  by  DeWitt  et  al.  [J.  Mofenson,  "Radar  Echoes 
from  the  Moon,"  Electronics,  Vol.  19,  No.  4,  92-98  (April  1946)].  These  experiments 
strongly  suggest  inadequacy  of  ionization  as  the  reason  for  absence  of  reflections. 

Qualitatively  propagation  in  this  range  strikingly  resembles  the  familiar  phenomena  of 
light.  Radio  "vision"  tends  to  be  limited  to  the  optical  line  of  sight,  though  diffraction 
actually  extends  coverage  considerably  beyond  obstructions,  such  as  hills  or  the  bulge  of 
the  earth,  except  for  extremely  high  radio  frequencies.  The  earth  (land  or  water)  for  many 
purposes  may  be  regarded  as  an  example  of  Lloyd's  mirror,  the  ground-located  receiver 
tending  to  be  in  the  first  dark  fringe  produced  by  reflection.  Refraction  tends  to  make  the 
distant  station  "visible,"  just  as  it  reveals  the  sun  a  few  minutes  before  sunrise,  and  it  pro- 
duces variations  and  anomalies  which  correspond  to  the  twinkling  of  stars  and  to  the 
mirage.  The  comparison  might  be  extended  to  other  phenomena. 

The  effect  of  regular  reflection  is  most  pronounced  in  the  meter  range,  though  even  in 
the  centimeter  range  cleared  land  or  water  may  make  a  good  "mirror"  for  glancing  inci- 
dence. In  such  cases  reflection  may  be  calculated  as  to  amplitude  and  phase  by  means  of 
standard  optical  formulas  provided  that  an  equivalent  conductivity  and  dielectric  constant 
are  known.  [P.  O.  Pederson,  The  Propagation  of  Radio  Waves,  Copenhagen,  1927;  C.  B. 
Feldman,  "The  Optical  Behavior  of  the  Ground  for  Short  Radio  Waves,"  Proc.  I.R.E., 
Vol.  21,  No.  6,  764  (June  1933);  Barfield,  J.  IJ8.E.,  Vol.  75,  No.  452,  214  (1934);  R.  L. 
Smith-Rose,  J.  I.E.E.,  Vol.  75,  No.  452,  221  (1934).]  As  has  already  been  implied,  though 
reflection  has  a  favorable  effect  with  long  waves  of  vertical  polarization,  causing  the  factor 
of  2  which  differentiates  eq.  (3)  from  (1),  in  this  range  it  is  commonly  unfavorable  owing 
to  the  phase  of  the  reflection  coefficient  caused  by  the  predominance  of  dielectric  currents 
in  the  ground  at  these  ultra-high  frequencies.  The  effect  of  reflection  need  not  be  unfavor- 
able, however,  if  one  or  both  of  the  terminals  is  located  at  a  sufficient  altitude.  At  high 
enough  altitudes  a  regular  succession  of  maxima  and  minima  is  encountered,  whose  posi- 
tion and  amplitudes  may  be  calculated  from  the  amplitude  and  phase  of  reflection. 

In  the  microwave  range  caution  is  necessary  in  applying  the  concept  of  reflection.  Here 
one  may  not  always  regard  the  earth  as  the  smooth  and  abrupt  boundary  between  materials 
described  simply  by  their  conductivities  and  specific  inductive  capacities.  Reflection  is 
not  always  specular  but  is  very  likely  to  be  diffuse,  as  the  success  of  radar  mapping  proves. 
One  should  not  ignore  even  here,  however,  the  strong  tendency  toward  specular  reflection 
that  glancing  incidence  imparts  to  the  scattering  from  a  rough  surface. 

Simple  plane-wave  reflection  theory  leads  to  a  useful  relationship  which  has  been  ob- 
served to  hold  with  fair  consistency  in  the  range  from  3  to  10  meters  and  even  in  the 
microwave  region  if  the  warning  in  the  last  paragraph  does  not  apply.  Over  level  land  or 
over  fresh  water  with  vertical  or  horizontal  antennas,  and  over  sea  water  with  horizontal 
antennas,  the  received  field  is: 


8  =  12ir  V5  volts  per  meter  (8) 

or        X 

the  radiated  power  P  being  in  watts,  and  d,  H\,  Hi,  and  X  in  meters.  HI  and  HZ  are  altitudes 
above  the  general  reflecting  area,  be  it  ocean,  valley  floor,  or  plain.  This  equation  is 
obtained  by  multiplying  the  free-space  field  of  eq.  (2)  by  ^irHiHz/hd,  a  procedure  justified 
if  HiH»_/\d  is  less  than  0.1  provided  that  the  reflecting  coefficient  of  the  ground  is  near 
unity  for  the  grazing  incidence  involved.  The  usefulness  of  eq.  (8)  is  not  limited  to  trans- 
mission over  plane  earth,  as  assumed  in  its  derivation,  but  roughly  applies  also  over 
spherical  earth  for  transmission  below  the  line  of  slight  when  3  meters  <  X  <  10  meters, 
3  meters  <  H  <  25  meters,  1  km  <  d  <  50  km.  In  this  extended  range  (B)  is  to  be 
regarded  as  an  empirical  formula  whose  range  of  validity  has  not  yet  been  determined. 
When  terminals  are  located  on  hills,  with  a  level  valley  between,  this  formula  needs  a 
correction  factor  due  to  ground  reflections  local  to  the  terminals. 


10-36  TRANSMISSION  cmCTJITS 

An  interesting  extension  of  eq.  (8),  valid  within  the  same  limitations,  is  obtained 
analogously  to  eq.  (13)  in  the  section  on  radio  antennas.  If  GT  and  GR  are  the  power  gains 
of  the  antennas  (in  terms  of  an  "isotropic  radiator"),  and  PT  and  PR  the  powers  trans- 
mitted and  received,  it  can  be  shown  that 


Note  that  power  received  is  independent  of  frequency.  If  the  antennas  are  short  doublets, 
G  =  1.5.  (K.  Bullington,  Proc.  I.R.E.,  1947.) 

With  vertical  antennas  over  sea  water,  but  otherwise  with  conditions  specified  above, 
an  approximate  inverse-square-of-distance  variation  has  been  found  up  to  30  or  40  km, 
with  the  important  difference  that  propagation  became  poorer  as  the  frequency  was  raised. 
This  is  in  accord  with  simple  optical  theory,  due  account  being  taken  of  the  electrical 
constants  of  sea  water.  Equation  (8)  therefore  does  not  apply  here,  though  it  does  for 
horizontal  antennas  over  sea  water. 

Since  diffraction  extends  the  range  beyond  obstacles  it  is  a  favorable  factor.  Ignoring 
diffraction  and  refraction  we  could  at  most  transmit  only_to  points  above  the  line  of  sight. 
The  range  would  then  be  limited  to  D  =  3500(V#i  +  Vjffa),  all  distances  being  in  meters. 
Within  this  range  eq.  (8)  is  applicable  unless  the  antennas  are  too  high,  with  qualifications 
already  mentioned.  Note  that  line  of  sight  does  not  at  all  guarantee  the  free-space  field 
strength. 

It  is  possible  to  make  instructive  calculations  of  the  fields  behind  obstructions,  such  as 
hills,  by  application  of  the  standard  mathematical  theory  of  diffraction.  Thus,  the  be- 
havior of  a  knife  edge  in  the  familiar  optical  example  becomes  a  guide  in  radio  transmission 
past  obstacles.  In  a  radio  problem,  in  order  to  obtain  a  reasonable  estimate  of  this  kind, 
the  effect  of  ground  reflection  at  the  transmitter  and  at  the  receiver  needs  to  be  taken  into 
account. 

If  the  index  of  refraction  of  air  is  calculated  as  a  function  of  height  from  average  meteor- 
ological conditions  [Humphreys,  Physics  of  the  Air,  McGraw-Hill  (1940),  p.  80],  the  gradual 
decrease  of  index  leads  to  a  "standard"  or  "normal"  condition  which  can  be  taken  into 
account  in  a  simple  manner.  Such  calculations  and  theoretical  considerations  indicate 
that,  if  the  topographic  cross-section  of  the  path  is  plotted  as  though  the  earth  had  a 
radius  4/3  times  its  actual  radius,  the  solution  of  the  corresponding  problem  assuming  a 
uniform  refractive  index  is  also  the  solution  for  the  actual  problem  including  the  effect  of 
refraction.  We  have  already  had  occasion  to  use  this  method  above  in  the  "spherical 
earth"  problem.  In  dealing  with  microwaves,  however,  this  simplication  will  be  misleading 
if  the  possibility  of  many  other  distributions  of  refractive  index  is  forgotten. 

In  connection  with  the  foregoing  topics,  reference  may  be  made  to  articles  in  Proc.  I.R.E., 
Vol.  21,  No.  3  (March  1933),  by  Jones  (p.  349);  Trevor  and  Carter  (p.  387);  Schelleng, 
Burrows,  and  Ferrell  (p.  427)  ;  and  Englund,  Crawford,  and  Mumford  (p.  464)  ;  also  Eng- 
hind,  Crawford,  and  Mumford,  B.S.T.J.,  Vol.  14,  No.  3,  369  (July  1935). 

At  shorter  wavelengths  refractive  variations  within  a  small  range  of  height  may  become 
important  because  the  wave  in  traveling  between  two  points  occupies  only  a  small  fraction 
of  a  kilometer  (that  is,  the  necessary  Fresnel  zones  are  now  included  in  a  small  transverse 
area).  It  results,  for  example,  that  such  waves  may  be  trapped  beneath  a  level  of  minimum 
refractive  index  and  may  travel  unusually  long  distances,  and  that  under  other  conditions 
they  may  unexpectedly  fail  over  short  ones. 

Refraction  theory  as  applied  to  extremely  short  waves  has  led  to  the  use  of  a  modified 
refractive  index  of  the  air,  M,  as  a  function  of  height,  h.  If  the  actual  index  as  ordinarily 
used  is  n(h]  and  the  radius  of  the  earth  is  a,  the  definition  M  -  10  ~*  =  n(h)  —  1  +  h/a 
leads  to  the  same  solution  with  an  assumed  fiat  earth  that  the  actual  index  leads  to  with 
curved  earth,  the  scale  factor  10""6  being  used  for  numerical  convenience. 

Just  as  short  waves  are  "bent  down"  by  the  ionosphere  because  its  refractive  index 
decreases  with  height  —  that  is,  the  phase  velocity  increases  with  height  —  so  in  the  tropo- 
sphere a  decrease  in  modified  index  will  tend  to  confine  microwaves  beneath  it.  Indeed, 
if  meteorological  conditions  are  such  as  to  give  a  maximum  index  at  a  certain  level  with 
progressively  smaller  values  above  and  below,  one  would  on  ray  theory  expect  the  ray  in 
its  horizontal  passage  to  undergo  consecutive  upward  and  downward  bendings  about  the 
level  of  maximum  index  (minimum  velocity)  .  Similarly  the  wave  might  be  confined  be- 
tween an  index  which  decreases  with  height  and  the  reflecting  floor  of  the  ocean  (or  perhaps 
of  land).  Such  phenomena  do  occur,  and  their  importance  is  that  they  may  lead  to  ab- 
normally high  or  abnormally  low  fields.  The  reason  for  the  strong  fields  is  that  cylindrical 
rather  than  spherical  expansion  causes  a  slower  decrease  with  distance,  i.e.,  increased 
range  horizontally.  The  phenomena  resulting  are  likely  to  be  complicated  and  variable, 
though  on  the  other  hand  such  "anomalies"  may  be  so  consistent  as  to  be  the  normal 


THE   SKY  WAVE  10-37 

condition.  ["9  cm  and  3  cm  Propagation  in  Low  Ocean  Ducts"  by  M.  Katzin,  R.  W. 
Bauchman,  and  Wm.  Binnian,  Dept.  of  Com.,  Office  of  the  Publication  Board,  Report 
PB  13747  (1945) ;  "Wave  Theoretical  Interpretation  of  Propagation  in  Low  Level  Ocean 
Ducts'*  by  C.  L.  Pekeris,  Dept.  of  Com.,  Office  of  the  Publication  Board,  Report  PB 
20228.]  Another  way  of  looking  at  these  ducts  is  to  use  wave  guide  concepts:  in  fact,  a 
duct  is  a  wave  guide  with  many  modes  of  propagation  which  are  excited  to  different  extents 
depending  on  the  elevation  of  the  transmitter  with  reference  to  that  of  the  duct.  The 
second  article  cited  discusses  the  subject  from  that  point  of  view.  For  a  summary  of  the 
war  work  dealing  with  the  various  aspects  of  this  problem  see  Radio  Wave  Propagation, 
Consolidated  Summary  Technical  Report  of  the  Committee  on  Propagation,  by  Burrows  and 
Attwood,  Academic  Press,  New  York. 

With  wavelengths  longer  than  10  cm  the  absorption  of  energy  from  the  wave  due  to 
the  atmosphere  itself  is  not  important  for  practical  purposes,  but  with  shorter  waves  at 
least  three  known  mechanisms  may  have  to  be  considered.  Water  in  the  liquid  or  solid 
phase  (e.g.,  rainfall)  is  one  of  these  that  can  become  very  serious  at  wavelengths  of  a 
centimeter  or  two.  The  loss  depends  on  the  total  water  per  unit  volume  and  on  the  size 
of  drop  or  particle  and  is  due  to  scattering.  Although  this  is  harmful  to  radio  transmission, 
the  phenomenon  is  being  utilized  by  meteorologists  in  the  detection  and  location  of  storm 
areas  by  means  of  radar.  In  the  vapor  phase,  water  has  its  longest  wave  resonance  at  1.3 
cm,  so  that  near  this  wavelength  high  absorption  is  to  be  expected  and  is  found  for  high 
humidities.  Oxygen  has  its  first  resonance  absorption  at  0.5  cm. 

Directional  properties  of  tropospheric  waves  have  been  studied  at  a  wavelength  3.25 
cm  by  W.  M.  Sharpless,  and  at  1.25  cm  by  A.  B.  Crawford  and  W,  M.  Sharpless,  Proc. 
I.R.E.,  Vol.  34,  No.  11,  837-848.  Deviations  in  azimuth  were  found  to  be  at  most  of  the 
order  of  0.1°.  Although  in  elevation  the  deviations  were  several  times  as  great  as  this, 
they  were  never  large.  Fading  about  the  free-space  field  strength  was  observed,  and  at 
times  the  field  exceeded  this  by  a  factor  of  4  (12  db).  At  times  there  were  multiple  waves 
coming  in  from  slightly  different  directions,  the  variations  of  which  produced  fading. 

22.   THE  SKY  WAVE 

THE  IONOSPHERE.  Whereas  for  most  radio  services  up  to  a  few  hundred  miles 
transmission  depends  on  the  direct  ground  wave,  for  all  long  distances  successful  transmis- 
sion depends  on  the  existence  of  a  "ceiling"  in  the  upper  atmosphere  which,  by  returning 
to  the  earth  the  outgoing  waves,  lays  down  a  useful  signal  at  distant  points  where  the 
ground  wave  is  negligible.  Even  the  longest  waves  used  in  radio  communication  depend 
on  such  "sky  waves"  for  distances  beyond  a  thousand  miles  or  two.  In  this  section  we 
shall  describe  present  views  of  this  ceiling,  the  common  name  for  which  is  the  "ionosphere," 
formerly  called  the  "Kennelly-Heaviside  layer."  The  first  suggestion  that  an  electrically 
conducting  region  exists  in  the  upper  atmosphere  was  made  by  Balfour  Stewart  to  explain 
variations  in  the  magnetic  field  of  the  earth.  Kennelly  and  Heaviside  were  the  first  to 
see  the  necessity  of  such  a  region  for  explaining  radio  transmission  phenomena. 

At  sea  level,  the  atmosphere  is  scarcely  conducting  at  all,  but  as  the  elevation  is  in- 
creased the  conductivity  increases  owing  to  an  increase  in  the  number  of  ions  in  unit 
volume.  This  ionic  density  increases  both  because  the  major  sources  of  ionization  are 
outside  the  earth  and  because,  at  the  lower  pressures  encountered,  the  ions  last  longer 
before  recombination  neutralizes  them  electrically.  Heavy  ions  (e.g.,  ionized  molecules) 
have  much  less  effect  on  radio  waves  than  electrons.  Electrons  are  known  to  exist  in 
sufficient  quantity  in  the  upper  atmosphere  to  produce  most  of  the  effect  observed,  but 
the  radio  effects  of  heavy  ions  have  not  as  yet  been  definitely  identified.  The  number  and 
distribution  of  these  ions  depend  on  various  factors,  including  altitude,  geographical  and 
geomagnetic  latitude,  local  time,  time  of  year,  and  solar  activity  (e.g.,  sunspots) .  The  varia- 
tion with  altitude  is  very  important.  It  has  been  found  that  the  increase  with  altitude  is 
not  at  all  uniform  and  simple,  but  that  there  are  regions  where  the  density  attains,  or 
tends  to  attain,  maximum  values.  Although  methods  thus  far  devised  are  not  suited  for 
studying  those  levels  above  the  maxima  where  (the  density  may  actually  decrease  with 
height,  it  is  probable  that  one  or  more  of  such  decreases  actually  occur.  These  levels  of 
mn.TriTm.im  (or  tendencies  thereto),  frequently  called  "layers,"  are  illustrated  in  Fig.  6. 
The  maximum  occurring  somewhat  above  100  km  is  known  as  the  E  layer,  and  those 
above  160  km  as  the  F  layers,  names  which  will  perhaps  be  superseded  when  the  mecha- 
nism of  their  production  is  explained.  It  is  fairly  definitely  known  that  ultraviolet  light 
from  the  sun  is  an  important  source  of  ionization,  particularly  in  the  E  region  and  in  one 
region  of  the  F  layer.  This  leads  to  the  large  differences  between  the  day  and  night 
behavior  of  radio  waves,  in  particular  to  phenomena  occurring  at  sunrise  and  sunset. 


10-38 


TRANSMISSION  CIRCUITS 


400 


Another  cause  of  ionization  is  particles  from  the  sun,  which  most  authorities  regard  as 
the  radiation  by  which  solar  disturbances  communicate  the  major  disturbing  effects  to  the 
earth.  These  are  in  particular  thought  to  be  the  cause  of  the  absorbing  "clouds"  which 
seem  to  form  in  polar  regions  beneath  the  E  layer  and  hinder  short-wave  transmission 
during  magnetic  storms.  The  distribution  of  ionization  changes  with  time,  and  Fig.  6  is 
to  be  regarded  merely  as  typical.  Tendencies  toward  maxima  come  and  go,  and  both 
the  E  and  F  regions  seem  to  be  composites  of  two  or  more  layers.  The  E  layer  is  the  most 
consistent  and  is  the  most  important  one  in  the  broadcast  band  (1500  kc  and  lower)  and 

at    lower    frequencies.      For 

50Oj 1 -j     these  lower  frequencies,  the 

waves  do  not  ordinarily  pene- 
trate higher  than  the  E  layer. 
For  the  lowest  frequencies  it  is 
not  known  whether  reflection 
occurs  at  the  E  layer  or  at 
some  lower  level,  there  being 
some  evidence  that  the  height 
is  80  or  90  km.  The  F  region 
is  of  most  importance  for  the 
short  waves,  particularly  over 
longer  distances,  but  the  E 
region  also  contributes  com- 
ponents to  the  signal,  which  is 
usually  very  complex. 

The  most  fruitful  method 
of  studying  these  regions  has 


E300 


~2QQf- 


100 


""«O 


F  Reglon 


-E  Region- 


Noon 


10° 


O  0  5xl06 

Electrons  per  Cubic  Centimeter 

3?iG.  6.  Concentration  of  Electrons  in  the  Upper  Atmosphere,  been  the  pulse  or  echo  method 
The  D  region  b  the  ionosphere  below  &0  km,  the  F  region  is  the  of  Breit  and  Tuve  [Phys.  Rev., 
part  above  160  km,  and  the  E  region,  is  the  part  between,  these  ,._.._  V  1  98  554  CloW^I  " 

which  a  very  short-wave  train 

of  a  given  frequency  is  radiated  upwards  and  the  times  required  for  various  reflections  to 
be  returned  are  obtained  with  an  oscillograph,  preferably  of  the  cathode-ray  type.  These 
times  may  be  converted  into  virtual  heights  of  the  reflecting  layer  by  assuming  the  pulse 
to  travel  with  the  velocity  of  light.  This  method  of  "radio  detection  and  ranging"  is  in 
fact  one  of  the  forerunners  of  radar.  Actually,  the  pulse  travels  with  a  slower  group 
velocity  than  this  while  in  the  ionized  region,  but  as  a  result  of  making  measurements  at 
several  frequencies  it  is  often  found  that  the  virtual  height  within  limits  is  nearly  independ- 
ent of  frequency,  and  for  such  frequencies  the  actual  cannot  be  very  different  from  the 
virtual  height.  Transmission  through  the  ionized  region  is  complicated  by  the  earth's 
magnetic  field,  which  makes  of  it  a  doubly  refracting  medium  in  which  the  wave  is  broken 
into  two  components  of  different  polarization  traveling  with  different  velocities.  It  is 
this  characteristic  which  has  led  to  the  identification  of  the  electron  as  the  active  ion.  The 
magnetic  field  leads  to  complications  in  practical  communication  by  causing  rotations  in 
the  plane  of  polarization,  leading  to  one  type  of  fading  (see  later  section  on  fading)  and  to 
errors  in  direction  finding  with  loop  aerials. 

It  is  natural  to  suppose  that  the  electrical  characteristics  of  the  ionized  region  are  linear, 
so  that  different  disturbances  may  be  superposed  without  interaction..  Evidence  has 
been  found,  however,  that  this  is  not  invariably  true  [B.  D.  H.  Tellegen,  Nature,  p.  840 
(June  10,  1933)3-  If  two  broadcasting  stations  of  high  power  operate  on  entirely  different 
wavelengths  and  are  separated  by  some  hundred  kilometers,  modulation  originally  im- 
pressed on  one  has  been  found  under  certain  conditions  to  have  been  transferred  to  the 
wave  of  the  second.  This  indicates  that  the  ionosphere  does  not  have  strictly  linear  char- 
acteristics. It  is  called  the  Luxemburg  effect. 

For  detailed  information  on  the  ionosphere,  and  for  bibliographic  references,  the  follow- 
ing may  be  of  interest:  E.  V.  Applet  on,  Inst.  E.E.  (London},  Vol.  7  (September  1932); 
Kirby,  Berkner,  and  Stuart,  Proc.  I.R.E.,  Vol.  22,  No.  4,  481  (April  1934) ;  Schafer  and 
Goodall,  Nature,  June  3  and  Sept.  30,  1933;  Bellinger,  Trans.  A.I.E.E.,  Supplement, 
Vol.  58,  SOS  (1939);  Darrow,  Bell  Sys.  Tech.  J.,  Vol.  19,  No.  3,  455  (July  1940). 

^  SKY-WAVE  PROPAGATION.  It  is  the  general  belief  that  waves  which  travel  long 
distances  do  so  by  means  of  multiple  reflections,  although  tbe  suggestion  has  been  made 
that  short  waves  (e.g.,  frequencies  above  3000  kc)  do  so  in  a  single  step.  Thus,  in  Fig.  7 
a  wave  is  conceived  to  travel  from  A  to  B  by  three  reflections  from  the  E  region  (3),  or 
by  fcwo  from  the  F  region  (2).  The  single-step  path  is  represented  by  curve  1.  If  at  each 
ioaospfeerie  reflection  double  refraction  due  to  the  earth's  magnetic  field  were  to  occur, 
for  tibe  two-reieetion  wave  not  one  but  four  (22)  components  might  be  found,  and  for  the 


THE  SKY  WAVE 


10-39 


three-reflection  mechanism,  eight.  Whether  the  complexity  is  thus  explained  or  not,  it  is 
a  fact  that  the  received  wave  is  frequently  very  complicated.  As  Fig.  7  suggests,  there  is 
wide  diversity  among  components  in  their  angles  of  elevation.  In  general,  however,  for 
all  these  paths,  the  received  energy  arrives  with  a  downward  component  of  velocity.  This 
is  of  great  importance  in  practice,  since  the  mode  of  transmission  places  important  direc- 
tional requirements  on  the  antennas  at  the  two  terminals.  In  the  horizontal  plane,  radio 
waves  as  a  general  thing  travel  along  the  great  circle  denned  by  the  locations  of  the 
terminals.  Consequently,  the  waves  on  arrival  are  usually  directed  approximately  along 


FIG.  7.     Sky-wave  Propagation  according  to  the  Ray  Theory 

the  true  bearing  of  the  transmitter,  regardless  of  frequency  range.  Small  variations  and 
differences  exist  in  azimuth,  with  occasional  large  ones.  See  Friis,  Feldman,  and  Sharpless, 
Proc.  I.R.E.,  Vol.  22,  No.  1,  47  (January  1934);  Friis  and  Feldman,  Proc.  I.R.E.,  Vol.  25, 
No.  7,841  (July  1937). 

LOW  FREQUENCIES  (LONG  "WAVES).  Among  the  chief  characteristics  here  are: 
(1)  at  a  given  frequency,  the  diurnal  variation  of  the  field  and  the  difference  between  day 
and  night  attenuation;  (2)  by  day,  the  greater  attenuation  at  the  higher  frequencies;  (3) 
by  night,  the  relatively  low  attenuation  and  the  relative  independence  of  attenuation  on 
frequency;  (4)  seasonal  variations;  (5)  propagation  substantially  along  the  great-circle 
path  and  the  departure  and  arrival  in  a  substantially  horizontal  direction;  and  (6)  the 
practical  absence  of  fading. 

Typical  diurnal  variations  of  field  strength  from  American  transatlantic  long-wave 
stations  as  received  in  England  are  shown  in  Fig.  8,  which  is  reproduced  from  Espensehied, 
Anderson,  and  Bailey,  Proc.  I.R.E.,  Vol.  14, 
No.  1,  7  (February  1926).    The  times  when 
the  path  is  entirely  in  daylight,  entirely  in 
darkness,  and  partially  in  each  are  shown  by 
the  shading  in  the  strip  at  the  bottom. 

Both  seasonal  variations  and  diurnal  varia- 
tions are  brought  about  by  the  changing  posi- 
tion of  the  path  relative  to  the  hemisphere 
illuminated  by  the  sun.  In  the  summer,  the 
duration  of  the  daylight  transmission  phe- 
nomena is  naturally  longer  than  in  the  winter 
on  account  of  the  longer  days.  It  is  this 
change  in  the  lengths  of  the  day  and  night 
periods  which  is  the  most  striking  feature  of 
the  seasonal  variation,  rather  than  any 
change  in  the  strength  of  signal.  When  the 
entire  path  is  illuminated  by  the  sun,  or  when 
the  entire  path  is  in  darkness,  the  character- 
istic day  or  night  phenomena  are  observed. 
When  the  path  is  half  illuminated,  half  dark- 
ened, a  characteristic  minimum  may  be  found 
in  the  diurnal  curve.  This  is  illustrated  in 
Fig.  8,  which  shows  a  pronounced  minimum 
occurring  near  sunset  in  the  57,000-cycle  curve. 

It  is  evident  that  the  field  to  be  obtained  at 
any  time  cannot  be  predicted  by  any  simple 
formula,  but  if  precision  is  not  required  it  is  possible  to  determine  its  order  of  magnitude 
for  the  night  and  for  the  day  condition.  Very  roughly,  the  midnight  field  in  long-distance 
transmission  has  an  average  of  the  order  one-fifth  the  inverse-distance  value.  By  day, 
the  fields  are  more  consistent  and  the  average  values  are  indicated  by  the  Austin-Cohen 
formula, 


G.M.T.  12 


10  12 


7     9    11 


135 
A.M. 


E.S.T.    7     9    1"!    1     3     5 
P.M. 

8.     Diurnal   Characteristics   of  Low  Fre- 
quencies in  Transatlantic  Propagation 


FIG. 


sin  u 


10-40 


TEANSMISSION  CIRCUITS 


where  So  is  given  by  (3)  or  (4),  #  in  radians  is  the  angle  subtended  at  the  center  of  earth 
by  the  path  D  is  in  kilometers,  and  /is  in  kilocycles  and  is  less  than  1000  kc.  These  con- 
stants were  suggested  by  Austin  in  1926  [Proc.  I.R.E.,  Vol.  14,  No.  3,  377  (June  1926)]  in 
order  to  make  the  formula  more  nearly  universal  for  daytime  transmission  than  the  original 
formula,  in  which  the  exponent  was  -87  X  lO^-D/0-5  A  convenient  form  is 


10* 


(10) 


in  which  P  is  radiated  power  in  kilowatts  and  E  is  in  microvolts  per  meter.  The  difficulty 
of  expressing  transmission  data  in  this  simple  and  usable  form  is  brought  out  by  the  fact 
that  in  one  part  of  the  frequency  range  covered,  namely  from  17  to  60  kc,  data  received 
in  transatlantic  transmission  are  better  represented  when  the  exponential  factor  is  modified 
to  s-4  x  iQ-W1'88  [Espenschied,  Anderson,  and  Bailey,  Proc.  I.R.E.,  Vol.  14,  No.  1,  7  (Feb- 
ruary 1926)]. 

Effects  accompanying  magnetic  storms,  and  secular  variation,   are  discussed  under 
"Solar  Disturbances"  in  article  23. 

INTERMEDIATE  FREQUENCY.     As  indicated  in  Fig.  2,  the  daytime  field  of  the 
sky  wave  in  this  range,  broadly  speaking,  is  attenuated  beyond  the  possibility  of  usefulness. 

Apparently  enough  ultra- 
violet light  from  the  sun 
penetrates  to  levels  of  the 
order  of  100  km  to  maintain 
an  absorbing  stratum  of 
ionization  in  spite  of  rela- 
tively rapid  recombination. 
Near  sunset,  however,  this 
cloud  disappears,  permit- 
ting a  considerable  reflection 
to  distant  points  on  the 
ground  during  the  night. 
There  is  still  absorption, 
and  it  is  variable  as  Fig.  9 
shows,  but  propagation  to 
long  distances  is  ordinarily 
possible.  (See  Standards  of 
Good  Engineering  Practice, 
loc.  cit.,  from  which  Fig.  9  is 
reproduced.)  Note  that 
~2000  2400  2800  this  figure  is  reasonably  con- 
sistent with  the  value  one- 
fifth  of  inverse-distance  field 
mentioned  in  the  preceding 
section  for  low  frequencies. 


AveUge  Uy  wave  field 

(corresponding  to  be  sfecond 

hodr  after  sunset  at  th 
•recordfng  sta.ion/ 


0.0002 
OjDQOl 


FIG.  9. 


400 


800 


12OO    1500 
Miles 


Night-time  Field  Strengths  from  250  to  2700  Mfles  (F.C.C. 
Data) 


To  a  first  approximation  night-time  transmission  in  this  range  is  independent  of  fre- 
quency. 

HIGH  FREQUENCIES  (SHORT  WAVES).  In  contrast  with  the  low-frequency  range 
discussed  above,  there  is  a  range  of  frequencies  above  approximately  3000  kc  m  which 
daylight  sky  wave  transmission  improves  with  increasing  frequency,  though  this  trend  is 
subsequently  reversed.  This  short-wave  range  is  limited  at  the  high-frequency  end  by 
the  inadequacy  of  electrons  per  unit  volume  of  the  ionosphere.  The  limiting  frequency 
in  the  daytime  is  not  very  different  from  30,000  kc  (10  meters).  Both  limits  are  variable, 
and  the  figures  given  are  somewhat  arbitrary.  Among  the  chief  characteristics  of  trans- 
mission in  this  range  are:  (1)  the  diurnal  variation  of  field  strength  and  the  prominence  of 
day-to-day  fluctuations;  (2)  the  greater  distances  of  transmission  obtained  with  the  higher 
frequencies,  especially  by  day;  (3)  the  "skip"  effect,  or  the  existence  of  a  region  about  the 
transmitter  in  whiefe  the  direct  wave  is  absent  owing  to  attenuation  of  the  ground  wave, 
and  the  sky  wave,  if  present  at  ail,  is  weak  and  erratic  owing  to  electron  limitation;  (4) 
habitual  fading,  sometimes  of  extreme  rapidity,  and  the  common  occurrence  of  selective 
fading;  (5)  the  necessity  in  most  cases  for  more  than  one  frequency  for  24-hour  service; 
(6)  the  great  reduction  of  field  strength  in  northern  and  southern  latitudes  concomitant 
wilfo  magnetic  storms  and,  by  contrast,  the  absence  of  a  pronounced  effect  in  equatorial 
regions,  and  other  phenomena  having  a  solar  origin;  (7)  a  secular  variation  following  the 
11-year  sunspot  cycle;  (S)  great-circle  transmission  and  a  wide  variety  of  angles  in  the 
vertical  plane. 


THE  SKY  WAVE 


10-41 


Diurnal  Variation.  Typical  diurnal  variations  are  shown  in  Fig.  10,  which  depicts  the 
changes  occurring  in  a  24-hour  interval  over  a  path  between  Deal,  N.  J.,  and  New  South- 
gate,  England,  the  radiated  power  being  1  kw.  The  curves  bring  out  the  advantages  of 
the  higher  frequencies  by  day  and  of  the  lower  by  night.  A  typical  curve  on  an  inter- 
mediate frequency  also  is  shown  (10.55 
me).  Day-to-day  variations  are  more 
pronounced  on  such  intermediate  frequen- 
cies than  for  either  higher  or  lower  fre- 
quencies. On  some  days  the  intermediate 
frequency  transmission  may  resemble  the 
lower  frequencies,  on  others  the  higher. 

Variation  with  Distance.  For  a  given 
distance,  the  transmission  conditions  de- 
pend on  the  geographical  latitude  of  the 
stations  and  to  some  extent  on  the  geo- 
magnetic latitudes,  the  difference  in  longi- 
tude, the  time  of  the  day,  the  time  of  the 


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year,  and  the  time  in  the  solar  cycle.    Fig- 

ure 2  will  serve  as  an  approximate  indica- 

tion of  the  frequencies  suitable  for  various 

distances  for  day  and  for  night.     The 

curves  were  based  largely  on  data  ob- 

tained during  the  years  1926  to  1930,  a 

period  which  included  a  sunspot  maxi-  5>  20 

mum.    In  general,  the  frequency  required 

is  higher  the  lower  the  latitude,  the  nearer 

the  time  to  noon  at  the  midpoint  of  the 

path,  and  the  nearer  to  the  secular  sun- 

spot  maximum.    In  the  winter  the  maxi- 

mum usable  (MUF)  frequency  for  mid- 

day is  greater  than  in  summer,  but  the 

lowest  usable  high  frequency  (LUHF)  is 

lower.    In  a  diurnal  curve,  the  character- 

istic day  and  night  conditions  are  obtained 

for  a  longer  period  the  more  uniform  the 

conditions  along  the  path.    Thus,  a  north- 

south  path  has  a  more  abrupt  change  from 

day  to  night  conditions  than  an  east-west 

path,  and  the  transition  condition  is  of 

shorter  duration.     This  transition  period 

is  relatively  difficult  because  no  single  fre- 

quency can  be  best  adapted  to  both  the 

day  and  night  portions  of  the  path.    Hence,  long  east-west  paths  tend  to  be  more  difficult 

than  north-south  paths.    Usually,  a  day  frequency  gives  a  weaker  field  by  day  than  a  night 

frequency  does  by  night.    The  variation  of  best  frequency  with  secular  magnetic  change 

is  exemplified  by  transatlantic  paths  such  as  that  from  New  York  to  London.    During 

the  sunspot  maximum  of  1930,  the 
best  daytime  frequency  was  about 
18T000  kc,  whereas,  during  the 
minimum  which  followed,  the  best 
was  under  15,000  kc.  (See  later 
discussion  under  "Maximum  and 
Minimum  Usable  High  Frequen- 
cies.*') 

The  Skip  Effect.    This  effect  is 
in  many  ways  analogous  to  the 
phenomenon  of  total  reflection  in 
optics.     A  light  wave    (Fig.    11) 
Phenomenon  of  Total   passing  from  one  medium  of  dielec- 


FIG.  10. 


Diurnal  Characteristics  of  High  Frequencies 
in  Transatlantic  Propagation 


<tfw/""//'tfw/w>twy/^^ 

\< Skip  Distance — *4 

Transmitter  ' 


FIG.  11.    The  Skip  Distance  as  £ 
Reflection 


trie  constant  n%  into  another  of 
smaller  dielectric  constant  ni  is 
subject  to  reflection  at  all  angles  if  the  change  in  dielectric  constant  is  abrupt,  and  to  total  re- 
flection for  angles  of  incidence  greater  than  ^  =  sin"1  ni/nz.  In  radio,  the  type  of  reflection 
resembling  total  reflection  is  believed  to  be  the  more  important,  since  the  change  in  dielec- 
tric constant  is  gradual.  For  large  angles  of  incidence  (incident  ray  approaching  the  hori- 


10-42  TRANSMISSION   CIRCUITS 

zontal)  the  wave  may  be  reflected  to  a  great  distance,  but  for  angles  less  than  the  critical 
angle  ^  (closer  approach  to  the  vertical)  the  wave  may  pass  through  without  substantial  re- 
flection. Although  the  radio  problem  is  more  complex  than  that  of  the  optical  illustration 
owing  to  the  gradualness  of  the  change  in  refractive  index  brought  about  by  the  gradual 
change  in  ionic  density,  in  both  cases  the  strong  wave  of  total  reflection  is  absent  at  the 
smaller  angles  of  incidence.  Since  the  index,  712,  of  air  at  sea  level  is  unity,  waves  incident  at 
the  ionosphere  at  an  angle  less  than  sin"1  ni  pass  through.  n\  is  to  be  taken  as  the  index  of 
the  ionized  region  at  the  altitude  of  maximum  ionic  density  (minimum  refractive  index), 
and,  since  it  is  a  function  of  frequency,  time,  and  other  factors,  the  skip  distance  is  a  func- 
tion of  the  same  factors.  Figure  2  shows  approximate  values  of  skip  distances,  obtained 
experimentally,  for  day  and  night  conditions  as  a  function  of  frequency.  [Taylor  and 
Hulburt,  Phys.  Rev.,  Vol.  27,  189  (February  1926).! 

Directions  of  Departure  and  Arrival.  For  a  single  wave  component,  this  direction  can 
be  expressed  by  two  angles.  One  is  that  in  the  horizontal  plane  and  can  be  given  either 
in  terms  of  true  north  or  as  deviations  from  the  great  circle  containing  the  transmitter. 
The  other  is  the  angle  in  the  vertical  plane  and  is  usually  given  as  the  angle  between  the 
ray  and  the  horizontal  plane. 

In  the  horizontal  plane,  the  direction  of  arrival  does  not  ordinarily  deviate  markedly 
from  the  great  circle,  deviations  greater  than  a  few  degrees  being  unusual. 

In  the  vertical  plane,  the  angle  with  the  horizontal  may  be  anywhere  in  the  range  from 
0°  to  90°,  depending  on  conditions  in  the  ionosphere,  distance  between  stations,  and  fre- 
quency. In  general,  as  constructions  such  as  that  of  Fig.  7  would  indicate,  the  angle  for 
short  paths  is  high.  For  long  distances,  the  angles  tend  to  be  small.  Thus,  in  transatlantic 
communication,  angles  from  10°  to  20°  are  common.  Angles  as  low  as  8°  and  as  high  as 
38°  have  been  measured.  The  average  seems  to  be  not  far  from  15°.  On  the  other  hand, 
signals  received  near  New  York  from  Buenos  Aires  commonly  arrive  at  vertical  angles  less 
than  5°.  In  the  case  of  the  transatlantic  paths,  a  great  deal  of  attention  has  been  given 
to  these  questions  by  Friis,  Feldman,  and  Sharpless,  Proc.  I.R.E.,  Vol.  22,  No.  1,  47 
(January  1934),  and  by  Feldman,  Proc.  I.R.E.,  Vol.  27,  No.  10,  635  (October  1939).  They 
found  that  the  directions  of  individual  wave  components  do  not  change  rapidly  or  capri- 
ciously; that  the  components  which  arrive  at  the  higher  angles  arrive  later  than  those  at 
the  lower,  qualitatively  as  might  be  expected  from  Fig.  7.  In  the  vertical  plane,  angular 
spreads  between  lowest  and  highest  component  have  been  found  at  times  to  be  smaller 
than  1°  and  at  other  times  as  large  as  20°. 

Polarization.  In  sky-wave  transmission  of  high  frequencies  the  composite  polarization 
of  the  received  wave  is  on  the  average  independent  of  that  transmitted.  The  direction 
of  the  electric  field  changes  in  a  random  manner  with  a  rapidity  which  is  connected  with 
that  of  fading. 

Echoes  of  Long  Delay.  Among  the  more  unusual  echoes  of  long  delay,  two  types 
are  of  particular  interest.  Under  certain  conditions,  "round-the-world  echoes"  can  be 
observed.  These  are  waves  having  a  delay  of  about  1/7  second,  which  travel  all  the  way 
along  a  path  which  probably  is  not  very  different  from  the  great  circle  separating  the  day 
and  the  night  hemispheres.  They  are  not  usually  observed  but  are  prevalent  at  certain 
times  of  the  year  for  a  given  pair  of  stations. 

A  second  type  of  echo  exhibits  delays  as  great  as  30  seconds.  This  extraordinary  retarda- 
tion may  be  due  to  extremely  low  group  velocities  in  the  ionosphere  or  to  waves  which 
travel  long  distances  outside  the  ionosphere  before  they  return.  They  are  as  rare  as  they 
are  mysterious. 

Maximum  and  Minimum  Usable  High  Frequencies.  Several  years  ago  the  National 
Bureau  of  Standards  began  a  systematic  study  and  publication  of  the  month-to-month 
variations  in  the  ionosphere  at  Washington,  D.  C.,  with  predictions  of  transmission  con- 
ditions to  be  expected.  During  the  war  this  work  was  developed  comprehensively.  From 
the  point  of  view  of  operation,  it  is  important  to  know  for  a  given  time  and  path  the 
maximum  usable  frequency  (MUF)  permitted  by  the  skip  phenomenon  and  the  lowest 
usaMe  high  frequency  (LUHF)  permitted  by  ionospheric  absorption.  Predictions  on  a 
worldwide  basis  are  available.  (See  publications  of  the  Central  Radio  Propagation  Lab- 
oratory, National  Bureau  of  Standards,  Washington  25,  D.  C.) 

23.  OBSTACLES  TO  TRANSMISSION 

ATMOSPHERIC  INTERFERENCE.  "Atmospherics"  or  "static"  are  electric  waves 
of  natural  origin  which  often  mar  radio  reception  or  make  it  impossible.  The  sounds 
prodiK^d  vary  from  the  crackling  of  extremely  short  impulses  called  "clicks,"  such  as 
may  be  produced  by  local  lightning  flashes,  to  the  steady  background  roar  called 


OBSTACLES  TO  TRANSMISSION 


10-43 


* 'grinders."     Hisslike  atmospherics  have  also  been  observed.    The  principal  characteris- 
tics of  atmospherics  may  be  listed  as  follows: 

1.  They  are  more  intense  in  summer  than  in  winter,  regardless  of  radio  frequency.    In 
the  northern  hemisphere  they  reach  their  maximum  in  July  or  August.    This  leads  to 
more  difficult  transmission  at  that  time. 

2.  They  have  a  diurnal  variation  in  intensity.    For  frequencies  below  about  10,000  kc 
the  night-time  intensity  is  greater  than  the  daytime.    The  difference  is  greatest  at  about 
1000  kc  in  the  vicinity  of  New  York  City,  being  perhaps  50  db,  and  is  very  small  at  15  kc. 
For  the  octave  above  10,000  kc  atmospherics  are  strongest  in  the  daytime  (see  Fig.  12). 


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Frequency 
FIG.  12.     Atmospheric  Noise  as  a  Function  of  Frequency  in  the  Vicinity  of  New  York  City 

The  diurnal  variation  has  a  definite  relation  to  the  diurnal  variation  in  the  transmission 
characteristics  of  the  frequency  used.  [C.  N.  Anderson,  Proc.  I.R.E.,  Vol.  21,  No.  10, 
1462  (October  1933).] 

3.  The  law  of  variation  of  amplitude  of  atmospherics  with  frequency  cannot  be  simply 
stated  and  is  in  fact  not  definitely  known.    Some  evidence  favors  an  inverse  first,  and  some 
an  inverse  second  power.    Figure  12  gives  one  estimate  of  the  frequency  distribution  [H.  K. 
Potter,  Proc.  I.R.E.,  Vol.  20,  No.  9,  1514  (September  1932)].    The  figure  is  for  reception 
in  the  northeastern  part  of  the  United  States;  elsewhere  the  intensity  and  its  distribution 
with  frequency  would  probably  be  somewhat  different.    By  day  the  amplitude  apparently 
follows  approximately  the  inverse  square  of  the  frequency  up  to  1000  kc.    At  higher  fre- 
quencies an  increase  having  a  maximum  at  about  15,000  kc  sets  in,  but  at  this  frequency 
noise  originating  in  the  receiver  is  likely  to  be  more  important.    The  night  curve  seems  to 
follow  the  inverse  first  power  up  to  about  10,000  kc.    The  advantage  of  low  frequencies 
due  to  better  transmission  is  evidently  reduced  by  the  greater  amount  of  atmospheric 
disturbance  encountered.     Likewise,  for  the  low  frequencies,  the  advantage  due  to  the 
fact  that  the  fields  at  night  are  stronger  than  day  fields  is  similarly  reduced  by  the  fact 
that  atmospherics  are  then  stronger. 

4.  Atmospherics  are  predominantly  of  tropical  origin.    Exceptions  are  those  on  inter- 
mediate frequencies  (500  to  3000  kc)  during  the  day,  and  atmospherics  on  all  ultra- 
ionospheric  waves.    These  are  of  comparatively  local  origin.    On  the  low  frequencies,  less 
atmospheric  interference  is  found  the  farther  the  receiver  is  removed  from  the  tropics. 
For  example,  they  are  less  in  Maine  than  near  New  York.     [Austin,  Proc.  I.R.E.,  June 
1926,  p.  373;  Espenschied,  Anderson  and  Bailey,  Proc.  I.R.E.,  Vol.  14,  No.  1,  7  (February 
1926).] 

5.  They  are  said  to  be  stronger  on  land  than  over  the  ocean.    [Austin,  Proc.  I.R.E,, 
Vol.  14,  No.  1,  133  (February  1926).] 

6.  Atmospherics  arrive  at  a  receiver  from  all  directions,  but  usually  certain  general 
directions  predominate  and  over  a  small  period  of  time  the  directivity  may^be  compara- 
tively sharp.    This  is  important  in  reception,  since  directional  discrimination  in  the  receiv- 
ing antenna  can  be  used  to  reduce  the  interference  without  reducing  the  desired  signal;  this 
is  not  possible  if  the  desired  signal  arrives  from  the  same  direction  as  the  atmospherics. 
When  there  is  a  variable  null  direction  in  the  antenna  polar  diagram,  the  improvement  ob- 
tainable is  very  considerable  unless  the  disturbances  arrive  through  a  wide  range  of  angles. 
[Espenschied,  Anderson,  and  Bailey,  Proc.  I.R.E.,  Vol.  14,  No.  1,  7  (February  1926).]  ^ 

This  directional  characteristic  is  due  to  the  existence  of  broad  centers  of  origin  which 
seem  to  coincide  with  thunderstorm  centers,  most  of  which  are  over  land.  Among  those  of 
greatest  importance  to  reception  in  the  United  States  are  Ecuador,  Brazil,  and  Central 
Africa  in  the  winter,  and  Mexico  and  Central  America  and  the  waters  between  there  and 


10-44  TRANSMISSION  CIRCUITS 

Florida,  Florida,  and  New  Mexico  in  the  summer.  The  sources  actually  causing  inter- 
ference will  depend  on  the  distance  range  of  the  frequency  used.  Thus,  frequencies  of  the 
order  of  1000  kc  will  have  only  local  static  by  day,  and  ultra-ionospheric  waves  will  never 
be  troubled  by  distant  sources.  At  night,  all  services  using  the  sky  wave  may  be  exposed 
to  distant  sources.  The  very  low  frequencies  are  exposed  to  distant  sources  at  all  times. 
(See  papers  by  Dean  and  by  Harper,  Proc.  I.R.E.,  July  1929,  pp.  1185  and  1214.) 

7.  Atmospherics  travel  along  the  earth  in  the  same  manner  as  signals  and  are  therefore 
subject  to  the  same  laws  of  attenuation.    Their  diurnal  variation  is  explained  principally 
by  this  fact. 

8.  Atmospherics  apparently  originate  in  discharges  with  sufficiently  abrupt  wave  fronts 
to  shock-excite  circuits  tuned  to  any  radio  frequency  lower  than  150  me  (or  some  higher 
limit;  see  next  paragraph).     Components  also  fall  in  the  audible  range  with  frequencies 
well  below  1  kc.     One  class  called  "tweeks"  has  a  limiting  frequency  between  1600  and 
1700  cycles,  suggesting  multiple  reflections  of  a  pulse  between  the  earth  and  a  conducting 
layer  90  km  above  it.    [Appleton,  Watson-Watt,  and  Herd,  Proc.  Roy.  Soc.,  A,  Vol.  Ill, 
165  (1926);  Burton  and  Boardman,  Proc.  I.R.E.,  Vol.  21,  No.  10,  1476  (October  1933)  .j 

9.  At  150  me  Schafer  and  Goodall  ['Teak  Field  Strength  of  Atmospherics  Due  to  Local 
Thunderstorms  at  150  Megacycles,"  Proc.  I.R.E.,  Vol.  27,  No.  3,  202-207  (March  1939)1 
found  (a)  that  the  peak  intensity  of  disturbances  varies  20  db  between  different  storms 
at  the  same  distance;  (5)  for  nearby  storms  the  inverse  distance  relation  is  a  good  approxi- 
mation for  the  calculation  of  the  variation  of  peak  disturbance  with  distance;  (c)  the  use 
of  high  instead  of  low  receiving  antennas  increases  the  signal-to-disturbance  ratio  almost 
directly  with  height  for  storms  within  10  miles;  (d)  the  durations  of  some  of  the  narrower 
peaks  in  any  particular  lightning  discharge  are  as  short  as  a  few  microseconds  or  shorter; 
(«)  the  maximum  equivalent  peak  field  for  a  storm  1  mile  away  was  about  0.015  volt  per 
meter  with  a  baad  width  of  1.5  megacycles.    Although  it  is  a  common  belief  that  atmos- 
pherics do  not  exist  at  microwaves,  actual  measurements  do  not  seem  to  have  been  reported. 

10.  Frequencies  in  the  band  near  20,000  kc  exhibit  distinctly  an  aural  difference  between 
atmospherics  from  local  and  from  distant  sources.    Atmospherics  from  local  sources  give 
the  * 'crash"  type;  those  from  distant  sources  ordinarily  give  a  fairly  steady  weak  back- 
ground.   The  difference  is  not  the  result  of  any  dissimilarity  in  mechanism  but  is  due  to 
the  skip  effect  which  excludes  disturbances  from  intermediate  distances.     The  direction 
of  arrival  of  the  steady  background  corresponds  with  that  observed  on  long  waves  (SW 
to  SE  near  New  York),  and  the  disturbance  is  heard  only  when  long-wave  static  is  very 
strong.    As  the  frequency  is  increased  into  the  ultra-ionospheric  range,  the  steady  back- 
ground disappears,  so  to  speak,  in  the  distance,  leaving  only  the  crashes  due  to  occasional 
local  storms.    A  weak  hisslike  disturbance  apparently  from  a  fixed  direction  in  space  and 
from  distances  beyond  the  confines  of  our  solar  system  has  been  observed  on  20,000  kc.    [K. 
Jansky,  Proc.  I.R.E.,  Vol.  20,  No.  12,  1920  (December  1932),  and  Vol.  21,  No.  10,  1387 
(October  1933).]     Jansky  pointed  out  that  this  noise  arrived  from  the  direction  of  the 
galactic  center  (Sagittarius),  and  Reber  [Asfrophysical  Journal,  Vol.  100,  279-287  (1944)], 
at  a  considerably  higher  frequency  which  permitted  high  directivity,  explored  the  region 
of  the  Milky  Way  and  found  noise  contours  corresponding  to  it.    Black-body  radiation 
from  the  sun  has  been  studied  by  G.  C.  Southworth  [J.  Franklin  Inst.,  Vol.  239,  No.  4, 
285  (April  1945)]  at  a  wavelength  of  3  cm;  it  may  be  looked  upon  as  resistance  noise,  the 
resistance  being  radiation  resistance  which  has  a  high  "temperature"  when  the  antenna 
points  at  the  sun.    An  enormous  increase  in  this  noise  has  been  reported  to  occur  at  times 
of  abnormal  solar  activity.    [Pawsey,  Payne-Scott,  and  McCready,  Nature,  Vol.  157,  No. 
39SO,  158  (Feb.  9,  1946),  and  Hey  and  Stratton,  Nature,  Vol.  157,  No.  3976,  47  (Jan.  12, 
1946).] 

11.  The  noise  level  differs  in  different  parts  of  the  sunspot  cycle.    At  high  frequencies 
there  is  a  direct  and  at  low  an  inverse  relation  to  sunspot  numbers. 

12.  Methods  of  combating  atmospherics  must  be  based  on  the  use  of  some  characteristic 
in  which  the  wave  of  the  signal  differs  from  that  of  the  disturbance.    The  most  important 
of  these  are  frequencyT  direction,  and  amplitude.    In  the  first,  selective  circuits  are  used 
which  suppress  current  of  frequencies  not  present,  or  necessary,  in  the  signal.    Even  with 
absolutely  ideal  selective  circuits,  an  irreducible  minimum  of  energy  will  pass  through  them, 
and  this  minimum  increases  linearly  with  the  frequency  range  necessary  for  signaling. 
(Carson,  B.S.T.J.,  April  1925.)    Directional  discrimination  has  already  been  discussed  in 
paragraph  6  above.    As  regards  amplitude,  the  most  obvious  procedure  is  to  increase  the 
effective  radiation  toward  the  receiving  terminal  by  increasing  either  the  power  capacity 
of  the  transmitting  set  or  the  effectiveness  of  the  transmitting  antenna.    Another  is  the 
tise  of  tfee  "compandor"  in  telephony,  by  which  the  low  amplitudes  are  raised  above  their 
masters!  vakse  wMle  passing  from  the  transmitter  (through  the  part  of  the  circuit  exposed 
to  staospbeoGs)  to  the  receiver,  the  normal  values  being  usually  restored  subsequently. 


OBSTACLES  TO   TRANSMISSION  10-45 

An  old  method  of  discrimination  against  atmospherics  of  high  amplitude  uses  opposed 
detectors,  equally  sensitive  at  high  amplitudes  but  sufficiently  different  at  low  amplitudes 
so  as  not  to  cancel  the  signal.  [Englund,  Proc.  I.R.E.,  Vol.  16,  No.  1,  27  (January  1928).] 

With  amplitude  modulation  (in  which  intelligence  is  carried  by  variations  of  the  ampli- 
tude from  its  average  value)  improved  signal-to-noise  ratio  can  thus  be  obtained  by  limiting 
the  range  of  frequencies  received  and  by  increasing  the  amplitude  variation  at  the  trans- 
mitter and  simultaneously  reducing  the  sensitivity  of  the  receiver  to  changes  in  amplitude. 
Analogously  with  frequency  modulation  (in  which  intelligence  is  carried  by  variations  of 
the  frequency  from  its  average  value),  the  interfering  effect  of  weak  noises  may  be  reduced 
by  limiting  the  range  of  amplitudes  received,  and  by  increasing  the  frequency  variation 
at  the  transmitter  while  simultaneously  reducing  the  sensitivity  of  the  receiver  to  change 
of  frequency. 

Some  forms,  such  as  ignition  noise,  occur  only  in  bursts  which  are  very  short  compared 
with  the  period  of  the  signal  being  transmitted  (e.g.,  in  telephony,  short  compared  with 
the  period  of  the  highest  audio  frequency).  It  is  advantageous  in  this  instance  even  with 
amplitude  modulation  to  widen  rather  than  narrow  the  frequency  band  of  the  receiver  and 
to  use  limiters;  this  widening,  it  is  true,  permits  more  noise  power  to  pass  into  the  final 
detector,  but  it  preserves  the  shortness  of  the  impulse  and  permits  the  limiter  to  chop 
off  the  peak,  leaving  a  pulse  no*  greater  than  that  of  the  signal  and  so  short  in  duration  as 
not  to  be  harmful. 

FADING.  In  its  most  general  sense  "fading"  means  a  reduction  of  the  signal  for  any 
cause,  including,  for  example,  the  slow  decrease  in  long-wave  signal  strength  called  the 
"sunset  minimum^"  which  may  last  an  hour  or  so.  Cornmonly  the  term  refers  to  the  more 
rapid  variations  encountered  with  medium  and  short  waves.  Probably  the  most  im- 
portant cause  of  fading  is  the  interference  of  wave  components  f oHowing  different  paths  in 
space,  combined  with  variations  of  phase  of  one  or  more  components  with  time.  The 
existence  of  the  different  components  can  be  due  to  multiple  reflections  or  to  double  refrac- 
tion due  to  the  earth's  magnetic  field,  and  perhaps  other  causes.  The  changing  with  time 
of  the  relative  phases  can  be  due  to  change  in  the  ionic  density  along  the  path  or  to  changes 
in  the  magnetic  field.  Pulse  experiments,  which  are  able  to  resolve  a  complex  signal  into 
many  components,  show  that,  with  the  best  resolution  possible,  the  components  themselves 
fade.  Fading  is  therefore  a  very  complex  phenomenon. 

An  important  example  of  fading  occurs  just  outside  the  service  area  of  a  high-power 
broadcasting  station.  With  increasing  distance  the  sky  wave  finally  becomes  appreciable 
compared  with  the  rapidly  attenuating  ground  wave.  Atmospheric  vagaries  make  the- 
relative  phase  of  the  components  vary  at  random,  causing  the  signal  level  to  fluctuate- 
It  may  be  further  complicated  by  the  reception  of  more  than  one  sky-wave  component* 
(See  article  24.) 

Rapidity  of  fading  of  ionospheric  waves  increases,  in  general,  with  frequency.  Below 
100  kc  the  phenomenon  is  scarcely  noticed;  such  changes  as  do  occur  commonly  require 
an  hour  or  so.  At  1000  kc  the  period  is  of  the  order  of  1  minute,  and  the  amplitude  range 
extreme.  At  10,000  kc  the  fading  rate  is  1  every  few  seconds.  Fading  due  to  interference 
between  two  or  more  components  should  show  this  characteristic,  for  a  fade  would  then 
occur  onee  for  every  change  of  1  wavelength  in  the  path  difference.  This  relation,  of  course, 
can  be  only  qualitative,  since  long  waves  and  short  waves  usually  travel  over  very  different 
paths,  are  frequently  not  used  at  the  same  time  of  day,  and  do  not  employ  the  same  number 
of  paths. 

A  distorted  frequency  characteristic  is  one  of  the  results  of  transmission  over  two  or  more 
interfering  paths.  Thus,  if  the  times  required  for  the  waves  to  travel  over  the  two  paths 
differ  by  r,  and  if  the  radio  frequency  is  slowly  varied,  consecutive  maxima  and  ininima 
will  be  found,  the  frequencies  of  the  maxima  differing  from  one  another  by  integral  multi- 
ples of  1/r.  If  r  =  0.0005  second,  a  typical  value,  the  maxima  will  be  separated  by  mul- 
tiples of  2000  cycles  per  second,  and  even  the  a-f  characteristic  may  be  seriously  affected. 
There  will  now  be  fading  provided  that  a  change  occurs  in  the  medium.  Being  a  function 
of  radio  frequency,  such  fading  is  called  "selective."  If  T  were  made  very  small,  the  fre- 
quencies of  the  maxima  would  differ  so  much  that  within  the  srrm-H  band  occupied  by  a 
telephone  channel  the  response  would  be  independent  of  frequency.  This  would  be  non- 
selective,  or  "general/*  fading.  The  selectiveness  of  fading  is  thus  associated  with  the 
time  difference  over  the  two  paths,  whereas  the  rate  of  fading  is  related  to  the  rate  of 
change  with  time  of  path  difference  measured  in  wavelengths.  It  is  common  to  have  more 
than  two  components  in  a  received  wave.  In  some  cases  the  wave  is  extremely  complex. 
[Potter,  Proc.  I.R.E.,  Vol.  18,  No.  4,  581  (April  1930).] 

As  shown  by  Bown,  Martin,  and  Potter  [Proc.  I.R^B.r  Vol.  14,  No.  1,  57  (February 
1926)}r  this  mechanism  can  produce  serious  distortion  in  a  speech  channel  if  the  instantane- 
ous frequency  of  the  transmitter  varies  during  the  audio  cycle. 


10-46 


TRANSMISSION  CIRCUITS 


The  change  in  field  strength  with  frequency  at  a  single  receiving  location  has  its  counter- 
part in  a  change  with  location  for  a  constant  frequency.  The  different  paths  differ  not 
only  in  length  but  also  hi  direction.  Most  important  perhaps  is  the  direction  in  the 
vertical  plane,  as  shown  in  Fig.  7,  but  the  directions  in  the  horizontal  plane  are  also  sig- 
nificant. It  is  these  directional  differences  which  cause  the  difference  in  signal  levels  at 
nearby  points,  for  the  two  waves  give  rise  to  a  set  of  interference  fringes.  Fading  does  not 
therefore  occur  simultaneously  at  nearby  points,  and  it  is  found  in  the  high-frequency 
range  that  points  separated  by  10  wavelengths  usually  fade  in  an  unrelated  manner, 
whereas  in  some  cases  the  separation  need  be  no  greater  than  2  or  3  wavelengths.  Advan- 
tage of  these  facts  is  taken  in  "diversity  systems"  of  reception,  which  employ  receivers 
operating  on  the  same  frequency  from  separate  receiving  antennas  at  different  locations 
or  different  polarizations,  or  on  different  frequencies  from  the  same  antenna.  [Beverage 
and  Peterson,  p.  531,  and  Peterson,  Beverage,  and  Moore,  Proc.  I.R.E.,  Vol.  19,  No.  4, 
562  (April  1931).} 

The  effect  of  fading  is  to  degrade  the  performance  of  a  circuit  used  in  communication. 
This  can  be  due  merely  to  reduction  of  field  during  the  fades,  which  leads  to  an  inadequate 
ratio  of  the  signal  strength  to  the  noise.  It  can  be  the  result  of  the  distorted  a-f  character- 
istic which  was  mentioned  above;  to  the  production  of  distortion  products,  as  for  example 
when  the  carrier  in  a  double  sideband  system  fades  out,  leaving  the  two  sidebands  to  beat 
with  each  other;  to  the  existence  of  fading  so  rapid  that  it  cannot  be  compensated  by  such 
devices  as  the  automatic  volume  control,  and  to  other  complications.  Probably  the  most 
usual  example  of  circuit  impairment  occurs  when  fading  and  noise  contribute  simultane- 
ously. All  such  effects  are  the  more  serious  the  higher  the  standard  set  for  the  circuit. 

SOLAR  DISTURBANCES.  Radio  transmission  is  one  of  the  terrestrial  phenomena 
which  may  be  correlated  with  solar  activity;  another  is  the  variations  which  occur  hi  the 

earth's  magnetic  field,  extreme 
fluctuations  having  been  given  the 
name  "magnetic  storms."  Except 
for  daylight  transmission  with  low 
frequencies  (e.g.,  60  kc)  the  effect 
of  unusual  solar  activity  is  an  ad- 
verse one.  By  day,  low  frequencies 
are  then  somewhat  aided,  but  at 
night  their  fields  are  considerably 
reduced.  The  most  marked  effect, 
however,  is  produced  on  the  high 
frequencies;  in  fact,  very  severe 
storms  may  completely  eliminate 
their  usefulness  over  some  paths. 
These  effects  are  illustrated  in  Fig. 
13,  which  shows  daytime  field 
strength  on  two  transatlantic  chan- 
nels, one  (full  line)  on  18,000  kc 
and  the  other  (heavy  dashes)  on  60 
kc,  together  with  the  horizontal 
component  of  the  earth's  magnetic 
field  (light  dashes).  [Anderson, 
Proc.  I.R.E.,  Vol.  17,  No.  9,  1528 


Strengths  In  db 

-  i  i  +  i 

no  01  O  01  o 

j 

<T»  0^  O»  Oi  vl  v|  u 
O>  ^1  00  ID  O  M  00 
O  O  O  O  O  O  -J 

H  Component  of  r> 
Earth's  Magnetic  Field 

i 

•••s^^ 

X 

> 

'—  -< 

>x 

\ 

/f 

£/- 

'   'N 

'  — 

r> 

v 

/ 

\ 

/ 

\ 

^ 

/ 

, 

^' 

•*""" 

T3         la 

E  -20 
-25 
~3Q 

l\ 

/I 

^^* 

/ 
/ 

r*-  " 

\ 

/ 

/ 

\ 

(/ 

54321012345678 
Days  Before      Storm             Days  After 

^  oTraasraissloo  on  18,34  MC  (IB  Meters)  Deal.  HJ.  to  Hew  Southgate,  England. 
<*— -sTransroission  on  60  Kc.(5000  Meters) Rocky  Point,  LI.  to  Cupar,  Scotland. 
»-  -- -tfodrontil  Component  of  Earth's  Magnetic  Field. 

FIG.  13.     Typical  Effects  Accompanying  Magnetic  Storms 


(September  1929).]  A  striking  demonstration  that  rapidly  moving  ionic  clouds  are  hurled 
into  the  ionosphere  during  magnetic  storms  has  been  given  by  Wells,  Watts,  and  George 
of  the  Carnegie  Institution  of  Washington  by  a  technique  of  rapidly  recording  the  reflec- 
tions received  over  an  extended  frequency  range. 

^The  solar  phenomena  include  sunspots,  prominences,  and  flocculi  which  may  be  observed 
with  a  telescope  or  spectrohelioscope.  The  disturbing  areas  rotate  with  the  sun  once  in 
every  27  days,  this  being  the  reason  for  the  ill-defined  "period"  of  27  days  in  the  terrestrial 
effects.  In  order  to  have  a  terrestrial  effect  it  seems  to  be  necessary  that  the  disturbed 
solar  area  have  a  certain  orientation  with  respect  to  the  earth.  The  27-day  period  in 
magnetic  and  radio  effects  forms  the  basis  for  a  method  of  predicting  future  disturbances; 
these  predictions,  though  not  entirely  reliable,  are  useful.  Another  periodicity  in  solar 
activity  is  the  secular  one,  having  a  cycle  of  about  11  years.  Since  the  last  minimum  and 
maximum  occurred  respectively  in  1944  and  1947,  it  appears  that  the  next  will  take  place 
about  1954  and  1958. 

Depending  on  the  severity  of  the  "storm,"  the  effect  may  last  one  or  several  days,  during 
which  communication  on  short  waves  is  erratic  and  difl&cult  and  sometimes  impossible. 
At  the  same  time,  the  aurora  may  be  visible,  earth  potentials  usually  rise  to  relatively 


RANGE   OF  RADIO  STATIONS,   BROADCAST  COVERAGE      10-47 


80 


545 
" 


|4Q 


\ 


Magi 
(Horiz 


\ 


\ 


\ 


90 

80 

270 

feo 
!«° 

§30 

<°20 
10 
0 


\ 


\ 


\ 


stic  Actlvi 
ntal  Ran 


Sig 


/\ 


high  values,  and  the  earth's  magnetic  field  may  be  seriously  disturbed.  Short-wave 
transmission  along  paths  through  equatorial  regions  are  scarcely  affected,  however;  it  is 
in  the  auroral  zone  that  the  effects  are  produced,  and,  apparently,  high-frequency  trans- 
mission over  any  path,  long  or  short,  which  requires  reflection  from  the  ionosphere  in  this 
zone,  is  adversely  affected.  At  such  times,  experiments  to  determine  virtual  heights  in 
these  zones  are  impossible  owing  to  the  total  absence  of  reflections.  [Appleton,  Naismith, 
and  Builder,  Nature,  Vol.  132,  No.  3331,  340  (Sept.  2,  1933).]  The  secular  period  in  this 
effect  has  already  been  mentioned.  Figure  14  illustrates  this  variation,  which  follows 
the  11-year  sunspot  cycle. 
[Austin,  Proc.  I.R.E.,  Vol.  90 
20,  No.  2,  280  (February 
1932).] 

Another  phenomenon  as- 
sociated with  the  sun  is  the 
radio  fade-out,  during  which 
all  sky  waves  except  those 
of  low  frequency  are  sud- 
denly weakened  or  obliter- 
ated over  the  earth's  sun-lit 
hemisphere.  It  has  been 
established  that  the  fade- 
out  is  coincident  with  a 
bright  solar  chromospheric 
eruption,  and  the  absence  of 
the  effect  at  night  and  its 
great  intensity  at  the  equa- 
tor indicate  ultraviolet  light 
rather  than  particles  as  the 
means  by  which  it  is  pro- 
duced. A  fade-out  may  last 
from  a  few  minutes  to  a  few 
hours.  It  is  evidently 
caused  by  an  unusually  high 
electronic  density  produced 
below  the  .S-region  which 
operates  by  absorption  due 
to  the  high  collision  fre- 
quency with  neutral  mole- 
cules at  that  level.  [J.  H. 
Bellinger,  Science,  Vol.  82, 
No.  2128,  351  (Oct.  11, 
1935) ;  L.  V.  Berkner,  Phys. 
Rev.,  Vol.  55,  No.  6,  536 
(March  15T  1939).] 

An  abnormality  known  as  sporadic  E  layer  reflections  is  of  some  importance  as  the  cause 
of  occasional  long-distance  transmission  at  frequencies  above  the  usual  ionospheric  limit, 
sometimes  as  high  as  60  me.  It  occurs  in  patches  rather  than  uniformly  over  the  E  layer 
and  is  not  well  understood.  [L.  V.  Berkner  and  H.  W.  Wells,  "Abnormal  lonization  of  the 
^-Region,"  Ter.  Mag.  and  Atmos.  Elec.,  Vol.  42,  No.  1,  73  (March  1937).] 


inspot  N 


\ 


Years 

FIG.  14.     The  "1 1-year"  Period  in  Sunspot  Activity  and  ItsjCorre- 
lation  with  Magnetic  Activity  _  and  Low-frequency  Radio  Trans- 
mission 


24.  RANGE  OF  RADIO  STATIONS  AND  BROADCAST  COVERAGE 

The  distance  over  which  communication  can  be  carried  on,  or  the  "range"  of  a  radio 
transmitting  station,  depends  on  so  many  changeable  phenomena  and  special  details  that 
the  term  is  usually  significant  only  as  an  order  of  magnitude  or  as  a  statistical  mean.  The 
range  depends  on  transmission  efficiency  and  noise  (e.g.,  atmospherics),  on  the  types  of 
apparatus  used  at  the  receiving  station,  and  on  the  standards  of  performance.  More 
meaning  attaches  to  the  range  of  a  ground  wave,  however,  because  of  its  steadiness  relative 
to  the  ionospheric  wave.  The  service  range  of  a  broadcast  station  may  be  set  by  noise 
due  to  atmospherics,  to  industrial  or  domestic  electrical  equipment,  or  to  unavoidable 
random  noise  arising  in  the  receiver  itself.  But  even  when  noise  is  negligible  it  may  be 
limited  by  fading  due  to  the  sky  wave's  being  appreciable  compared  with  the  ground  wave. 

Considering  noise  as  the  limiting  factor,  we  may  start  with  representative  noise  data 
such  as  those  given  in  Fig.  15.  This  figure  gives  approximate  noise  values  (atmospherics, 


10-48 


TRANSMISSION  CIRCUITS 


set  noise,  etc.)  which  we  assume  as  typical  for  medium-frequency  broadcast  reception  in 
northeastern  United  States  in  the  summer.  For  reception  the  signal  field  must  be  greater 
than  the  noise  by  certain  values  which  depend  on  the  grade  of  service  desired.  This  leads 
to  a  required  signal  field,  and  the  distance  at  which  it  is  obtained  under  certain  conditions 
can  be  obtained  by  reference  to  Fig.  3.  For  example,  to  find  the  summer  night  range  of 
the  ground  wave  of  a  100-kw  transmitter  operating  on  1500  kc,  we  have: 

Summer  midnight  noise  on  1500  kc,  Fig,  15 0.035  mv/meter 

Signal-to-noise  ratio  assumed ' 100 

Field  required  with  100  kw  radiated^ 3.5    mv/meter 

Field  with  1  kw  radiated  (3.5A/100) • 0.35    mv/meter 

Distance  giving  0.35  mv/meter  (51  db  above  1  mv/meter)  with  1  kw, 

<r  =  2  X  10~13,  Fig.  3 53  miles 

Figures  such  as  these,  which  are  based  on  transmission  data  applying  in  the  ease  of  level 
terrain,  cannot  in  general  apply  if  there  are  large  obstacles  or  other  irregularities  in  the 

path.   A  striking  example  of 
1,0  c  — — 


0.1 


I 


0.01 


:  Noise  Level  vs  Frequency .; 


'   ftjninj 


this  was  described  by  Bown 
and  Gillette,  who  found 
that  sections  of  New  York 
City  in  which  there  are  large 
numbers  of  unusually  tall 
buildings  cast  "shadows" 
for  several  miles  [Proc. 
I.R.E.,  Vol.  12,  No.  4,  395 
(August  1924)].  Dense 
areas  of  small  buildings  also 
reduce  the  range.  There 
are  other  factors  than  noise, 
such  as  fading  and  interfer- 
ence from  other  stations, 
that  complicate  this  prob- 
lem and  require  consider- 
able experience  to  assess. 
Field  strength  surveys  of 
the  actual  territory  to  be 
covered  are  frequently  re- 
quired. [Standards  of  Good 
Engineering  Practice, 
F.C.C.,  loc.  cit.;R.  F.Guy, 
loc.  cit.;  "An  Analysis  of 
Continuous  Records  of  Field 
Intensity  at  Broadcast  Fre- 
quencies," Norton,  Kirby, 
and  Lester,  Proc.  I.R.E., 
Vol.  23,  No.  10, 1183  (Octo- 
ber 1935);  "On  the  Use  of 
Field  Intensity  Measure- 
ments for  the  Determina- 
tion of  Broadcast  Station 
Coverage,"  Jansky  and 
Bailey,  Proc.  I.R.E.,  Vol. 
20,  No.  1,  62  (January 
1932).] 

Another  limitation  in  the 
service  area  of  a  broadcast 
station  in  the  evening  is  the  existence  of  fading  beyond  a  certain  distance.  This  limiting 
distance  depends  on  frequency  and  earth  conductivity,  and  in  general  it  is  greater  the 
lower  the  frequency  and  the  higher  the  conductivity.  Fading  first  becomes  serious  when 
the  sky  wave  becomes  appreciable  compared  with  the  ground  wave.  Figure  16  illustrates 
as  a  function  of  frequency,  how  the  distance  range  of  broadcast  stations  depends  on 
electrical  noise  and  fading,  an  earth  conductivity  of  10~13  being  assumed.  [Report  of  Com- 
mittee on  Radio  Propagation  Data,  Proc.  /.£.#.,  Vol.  21,  No.  10,  1430  (October  1933)  ] 


i  o.ooi 


I 


0.0001 


0.00001 


8  S 


800000000 
oooooooo 
o    O    1-1    CM    to     ^rtnto     pv 


Frequency  -  Kilocycles 
FIG.  15.    Typical  Noise  Data,  Broadcast  Reception 


TRANSMISSION-LINE   CONSTRUCTION 


10-49 


Midnight 


1000, 


Mtfday* 
Ground  Wave  Transmission  Only 


Kilocycles 


8     8 

CNJ  «* 

Variations  with  Changes  in  Power 


800 
600 
400 

52200 
<o 

lioo 

o    80 
5    60 
40 

20 

10 
< 

_i 

b 

^N 

•^, 

ai 

N< 

ie- 
w 

Su 
Yo 
lo 

mr 

rk- 

id, 

•>-. 

ne 
Su 
i-S 

r,  - 

mn 
ur 

F 
ner 
•im 

or 

da 

-W 

nt 

er 

s. 

—  — 

i*! 

•••. 

••^ 

— 

== 

Power  1  Kw 
%  Ratio^i 
Minimum  Noise  .01MVw 

=    888888888 

Kilocycle 


Variations  with  Changes  in  Noise  Level         Kilocycles 


auu 
600 
400 

2200 
£100 

-*. 

•«v 

^: 

•**» 

-». 

s, 

x 

v 

^ 

--. 

•-v 

••». 
••», 

-^. 

-«. 

iti 

>  1 

k 

^    60 
40 

20 

zr4' 

YI 

Min 

Power  1  Kw 
mum  Noise  .01M>w 

1 

8888888 

--         CO          O         CM         ^         u>        w 

Kilocycles Kilocycles 

Variations  with  Changes  in  Signal-to-Noise  Ratio 

*Assuming  most  favorable  daytime  conditions  and 
limiting  background  noise  of  .01,  in^millivolts  per 
meter  throughout  the  frequency  ranga. 

FIG.  16.    Useful  Hange  of  Broadcast  Stations  under  Different  Conditions  of  Power,  Noise  Level, 
and  Permissible  Signal-to-noise  Ratio.     Central  United  States  and  Europe. 

MECHANICAL  FEATURES  OF  TRANSMISSION  LINES 

By  John  D.  Taylor 


25.  TRANSMISSION-LINE  CONSTRUCTION 

POLE  LINES  are  employed  in  aerial  communication  construction  to  support  open 
wires  and  cables  used  in  toll  and  exchange  plant.  The  supporting  structures  are  generally 
of  wood  but,  for  special  requirements  or  where  pole  timber  is  not  obtainable,  may  be  of 
steel  or  'other  materials. 

The  use  of  wood  poles  generally  throughout  the  United  States  and  other  countries  is 
due  principally  to  (1)  availability,  (2)  economical  type  of  construction,  (3)  ease  of  handling 
and  maintaining,  and  (4)  relatively  long  life. 

The  design  of  wood-pole  lines  is  based  primarily  on  (1)  type  of  communication  plant 
(toll  or  exchange)  to  be  supported,  (2)  load  to  be  carried,  and  (3)  location  and  exposure  to 
weather.  The  poles  must  be  of  sufficient  strength  (allowing  for  ground  decay  and  econom- 


10-50 


TRANSMISSION  CIRCUITS 


ical  life)  and  height  (allowing  for  ultimate  loads  and  required  clearances)  to  meet  the 
requirements  for  this  type  of  construction  in  the  most  economical  manner. 

In  exchange  plant,  initial  pole-line  routes  largely  determine  the  routes  of  distribution 
for  succeeding  types  of  construction,  while  for  toll  plant  the  large  initial  costs  involved  in 
building  a  toll  pole  line  usually  require  its  maintenance  on  the  selected  route,  at  least  for 
its  economical  life  or  until  other  considerations  such  as  right-of-way,  growth,  new  develop- 
ments or  deterioration  necessitate  its  removal  or  replacement.  In  planning  new  pole  lines 
or  rearrangements,  it  is  important  to  so  advise  other  pole-using  companies  in  the  area 
involved,  in  order  that  the  plans  of  all  the  companies  may  be  in  coordination  at  all  times. 

The  selection  of  the  pole  route  usually  entails  advance  surveys,  acquiring  the  necessary 
right-of-way,  and  other  considerations,  which  will  provide  the  required  pole  line  econom- 
ically. Due  regard  must  be  given  to  the  future  adaptability  and  relation  of  the  pole  line 
with  respect  to  the  telephone  system,  serving  the  area,  as  a  whole.  Toll  pole  lines  usually 
take  the  most  direct,  practicable  route  between  terminating  points,  avoiding  small  towns, 
trees,  and  hazardous  conditions  as  far  as  possible.  Toll  points,  off  the  main  toll  route,  are 
reached  by  branch  (spur)  lines.  Along  highways,  one  side  should  be  occupied  throughout, 
as  consistently  as  conditions  permit,  avoiding  unnecessary  road  crossings  and  leaving  the 
other  side  of  the  highway  for  other  wire-using  companies. 

Public  right-of-way  is  less  expensive  initially,  but  private  right-of-way  for  a  particular 
section  of  pole  line  may  result  in  lower  annual  charges  and  ultimate  costs  and  most  cer- 
tainly will  add  to  the  permanence  and  safety  of  the  line. 

Joint  use  is  generally  desirable  and  economical  in  urban  areas,  rather  than  using 
separate  pole  lines  for  power  and  telephone  facility  distribution.  The  power  circuit 
vohages  in  cities  are  usually  low  (not  over  5000  volts),  and  the  telephone  equipment  and 
subscribers  axe  adequately  protected  in  case  of  contact  between  the  power  and  telephone 
circuits. 

Joint  use  with  rural  or  toll  open-wire  circuits  is  not,  as  a  rule,  desirable,  because  of  the 
generally  higher-power  circuit  voltages  and  of  possible  hazards  to  life  and  property  from 
contacts.  Consideration  may  be  given,  however,  to  such  joint  use  in  any  particular  case, 
and  future  developments  may  indicate  its  desirability  for  rural  construction. 

The  selection  of  poles  required  for  any  particular  pole  line  is  based  mainly  on  (1)  the 
number  of  telephone  or  other  aerial  wires  and  cables  to  provide  facilities  over  the  expected 
service  life  of  the  poles,  (2)  the  importance  of  these  facilities,  (3)  the  pole  strength  required 
to  carry  the  initial  and  ultimate  wire  and  cable  loads  under  the  weather  conditions  ex- 
pected in  the  locality  involved,  and  (4)  governmental  regulations. 

Some  companies  have  established  classifications  for  open-wire  and  cable  pole  lines,  in 
accordance  with  the  service  value  of  the  line  (relative  importance  and  number  of  messages 
carried  by  the  circuits  on  the  line).  Figure  1  shows  the  classifications  used  by  the  Bell 
System  and  the  relative  strengths  assigned  based  on  the  System's  experience.  Lower 
percentages  of  ultimate  stress  are  required  for  railroad,  power,  or  similar  crossings. 


Classification 
of  Line 

Type  of  Service 

Message- 
miles 
per  day 

Relative 
Strength 
Levels 

Maximum  Percentage  of 
Ultimate  Fiber  Stress 
under  Transverse  Loading 

For  New 
Poles 

At  Replace- 
ment 

A 

Toll  open-vrire  and  cable  of 
high  service  value 

Over  30,000 

100 

45 

67 

B 

Toll  open-wire  (average  serv- 
ice value)  and  toll  cable 
(not  class  A) 

5,000  to 
30,000 

67 

67 

100 

C 

Exchange  open-  wire  (over  10 
"wires),  all  exchange  cable.  .  . 

50 

89 

133 

Toll  open-wire  of  low  service 
value 

Less  than 
5,000 

50 

89 

133 

R 

Exchange  open-wire  (not  over 
1  0  wires)  

33 

133 

200 

J 

Joint  power  and  telephone 
lines  

88 

50 

75 

FIG.  1.    Classification  and  Strength  Requirements  of  Pole  Lines,  Bell  System  Practices 


TRANSMISSION-LINE  CONSTRUCTION 


10-51 


In  connection  with  the  preparation  of  the  National  Electrical  Safety  Code  (N.E.S.  Code) , 
National  Bureau  of  Standards  Handbook  H32,  studies  were  made  to  determine  the  fre- 
quency, severity,  and  effects  of  ice  and  wind  storms  throughout  the  country.  On  the 
basis  of  these  studies,  three  general  loading  areas,  heavy,  medium,  and  light,  were  estab- 
lished for  the  United  States,  as  shown  in  Fig.  2.  For  the  same  classification  of  line  and 
other  structural  condi- 
tions, a  heavier  class  of 
pole  is  required  (particu- 
larly for  average  or  greater 
loads)  in  the  heavy  than  in 
the  medium  or  light  load- 
ing areas,  and  likewise  in 
the  medium  as  compared 
to  the  light  loading  area. 

Basic  conductor  load- 
ings have  been  assigned 
for  the  three  loading  areas, 
in  order  to  derive  pole 
loadings,  considered  ap- 
propriate for  these  areas, 
and  with  the  various  pole- 
line  classifications  and 
other  data,  to  arrive  at  the 
class  of  pole  required  for 
any  given  line.  Figure  3 
shows  these  assumed  load- 
ings and  associated  constants,  which  latter,  when  added  to  the  resultant  of  the  vertical 
and  horizontal  loads,  will  result  in  effective  conductor  loadings,  substantially  the  same 
for  the  Fifth  Edition  as  for  the  Fourth  Edition  of  the  N.E.S.  Code.  Thus,  it  has  been 
possible  to  avoid  lowering  past  overall  effective  standards  for  pole-line  strength  require- 
ments, and  at  the  same  time  to  reduce,  in  the  Fifth  Edition,  the  transverse  loadings  on  the 
pole  line  to  permit  the  use  of  allowable  stress  values  more  nearly  representative  of  general 
engineering  practices. 

Assumed  Vertical  and  Transverse  Loadings 


G.  2.    Storm  Loading  Map  of  the  United  States  (from  N.E.S.  Code, 
Fifth  Edition,  Natl.  Bu.  Stds.  Handbook  H8£) 


Radial  Thick- 

Horizontal Wind 

Storm 

ness  *  of  Ice 

Pressure  at  Right 

Loading 

Symbol 

Coating  on 

Angles  to  the  Line, 

Area 

Conductors  and 

lb/sq  ft  of 

Messengers,  in. 

projected  area 

Heavy 

H 

0.50 

4 

Medium.  ...... 

M 

0.25 

4 

Light  

L 

None 

9 

*  Note:  In  computing  transverse  loading  on  poles  and  towers,  ice  coating  on  these  structures  is 

ignored. 


Constants  for  Various  Types  of  Conductors  to  Be  Added  to  the  Resultants  of  the  Loadings 
Shown  in  above  Table,  pounds  per  foot 


Storm 
Loading  Area 

Sym- 
bol 

Tem- 
pera- 
ture, 
degF 

Bare  Copper,  Steel, 
Copper  Alloy, 
Copper-covered 
Steel,  and  Combi- 
nations Thereof 

Bare  Aluminum 
with  or  without 
Steel 
Reinforcement 

Weather  Proof 
and  Similar 
Covered 
Conductors 
(AH  Materials) 

Cable 
with 
Messenger 

H 

0 

0.29 

0.31 

0.31 

0.6 

M 

+  15 

.19 

.22 

.22 

.4 

Light 

L 

+  30 

.05 

.05 

.05 

.2 

Note:  For  telephone  wires  it  is  usually  assumed  that 

P  =  0.003  V2 

where  P  —  horizontal  wind  pressure  in  pounds  per  square  foot  of  projected  area. 
V  —  actual  wind  velocity  in  miles  per  hour. 

FIG.  3.     Assumed  Vertical  and  Transverse  Loadings  and  Associated  Constants  (from  N.E.S.  Code, 
Fifth  Edition,  National  Bureau  of  Standards  Handbook  HSS) 


10-52 


TRANSMISSION  CIRCUITS 


Since  pole  lines  carry  various  types  of  communication  facilities  and  frequently  (on  joint 
lines)  power  conductors  and  equipment  as  well,  it  is  necessary,  in  determining  pole-line 
loads,  to  equate  the  various  attachments  to  a  common  basis.  Figure  4  gives  wire  equiva- 
lent data  for  the  three  loading  areas  in  terms  of  effective  104  (mil  diameter)  telephone  wires 
and  of  effective  No.  4  covered  power  wires.  In  heavy  and  medium  loading  areas  only, 


Attachment 

Telephone  Wire  Base 
(Effective 
104  Tel.  Wires) 

Power  Wire  Base 
(Effective  No.  4 
Covered  Power  Wires) 

Storm  Loading  Area 

Storm  Loading  Area 

Heavy 

Medium 

Light 

Heavy 

Medium 

Light 

Communication  plant 
Bare  open  wire,  109,  104,  or  smaller,  per  wire 
Bare  open  wire,  128,  134,  or  larger,  per  wire. 
Covered  paired   wire,   per  pair,   or  covered 

1 

1 

1 
2 
4 
5 
6 
7 

1 

1 
1 
1 

8 

4 

1 

1 

1 
2 
5 
6 
7 
8 

1 

1 

1 

14 

7 

I 
1.3 

2.5 
5 
15 
22 
30 
35 

4 

6 
4 
2 

50 

25 

0.8 
0.8 

0.8 
2 
3 
4 
5 
6 

1 

1 
1 

3 

0.7 
0.7 

0.7 
2 
3 
4 
5 
6 

1 

1 
1 
0.5 

5 

0.3 
0.4 

0.7 
2 
4 
6 
8 
10 

1 

2 

1 
0.5 

8 

Cable  and  1  0  000  Ib  (3/8  i11-)  strand 

Cable  and  1  6,000  Ib  (7/16  in  )  strand        .    ... 

Cable  and  25  000  Ib  (1/2  ^  )  strand  

Cable  terminal,  "B"  or  "BB"  type,  202  pair 
and  less  or  "BD"  type,  all  sizes 
Cable  terminal,  "B"  or  "BB"  type,  more  than 
202  pair                                

Cable  loading  pot                -  -  -    -  -  • 

Service  drops  'per  unbalanced  drop 

Clothes-lines  on  Class  C  line  poles,  per  unbal- 

GLolhee-lines  on  Class  J  line  poles,  per  unbal- 

Power  plant 
Covered  wire,  No.  8  AWG  (approx.  0.26  La. 
o  d  )  or  smaller  per  wire 

1.1 
1.2 
1.3 
1.5 
2 
3 
3 
6 

6 
12 
18 

2 
3 

5 
1 
1 

1 
I 

I 

1.3 
1.4 
1.5 
1.9 
3 
4 
5 
7 

10 
20 
30 

2 
3 

6 
1 
2 

2 

1 

1 

2.5 
3.1 
3.7 
6.3 
11 
15 
20 
30 

25 
50 
75 

8 
9 

19 
6 
9 

4 
3 

3 

0.9 
1.0 
1 
1.2 
2 
3 
3 
5 

5 

10 

15 

2 
3 

4 
1 

1 

1 
1 

1 

0.9 
1.0 
1 
1.3 
2 
3 
4 
5 

7 

14 
21 

2 
2 
4 
2 
2 

1 

0.7 
0.9 
1 
1.7 
3 
4 
6 
8 

7 

14 
21 

2 
3 

5 

2 
3 

1 

1 

1 

Covered  wire,  No.  6  AWG  (appro*.  0.32  in. 
o  d.)  per  wire      

Covered  wire,  No.  4  AWG  (approx.  0.38  in. 
o  d  )  per  wire                  ... 

Covered  wire,  No.  0000,  AWG  (approx.  0.65 
in.  o  d.)   per  wire  

Covered  wire,  500,000  circ  mils  (approx.  1.11 

Covered  wire,    1,000,000  circ  mils   (approx. 
1  53  in  o  d.)   per  wire            .             . 

Covered  wire,    2,000,000  circ  mils   (approx. 
215  in.  o.d.)  or  larger,  per  wire 

Power  cable   on  strand    (approx.   diam.   of 
cable  2  56  in  or  less)  .  .  . 

Suspension  wire  extend- 
ing   transversely   be-    One  contact  wire 
tween  two  pole  lines    Two  contact  wires 
and  supporting  trolley    Four  contact  wires 
contact  wires,  per  line 
Bracket  and  one  trolley  contact  wire  on  one 
side  of  pole  line 

Brackets  and  two  trolley  contact  wires,  one 
on  each  side  of  pole  line       ...      -      .  .    ... 

Bracket  and  two  trolley  contact  wires,  over 
tracks  on  same  side  of  pole  line 

Tr&Tvgforrnpns  37  l/o  fc-u-a  or  less 

Transformers,  over  37  !/•>  kva  

Transverse  clearance  attachment  for  service 
drop  above  telephone  attachments,  per  wire 
Service  drops,  per  unbalanced  drop  wire  
Street   lamp   supported   by   mast-arm    (not 
bracket)  ,  

Table  of  Wire  Equivalents  for  Pole-line  Loading  Calculations  (from  NJS.S.  Code,  Fifth  Edition, 
Notional  Bureau  of  Standards  Handbook  HS%} 


TRANSMISSION-LINE   CONSTRUCTION  10-53 

when  the  actual  number  of  wires  on  crossarms  is  more  than  10,  shielding  reduces  the  effec- 
tive number  of  wires  to  67  per  cent  of  the  actual  number.  In  light  loading  areas,  loading 
is  assumed  on  actual  wires. 

The  selection  of  new  poles  for  a  given  pole  line  requires  that  the  following  factors  be 
known  or  assumed: 

(a)  Classification  of  pole  line  (determines  the  relative  importance  of  the  line), 

(6)   Loading  area  (determines  the  basic  loadings). 

(c)   Fiber  strength  of  timber  to  Table  1 

be  used  (see  Table  1). 


Type  of  Timber 


Fiber  Strength, 
Ib/sq  in. 


Northern  white  cedar 

"Western  red  cedar 

Creosoted  southern  pine  and  Douglas  fir. . 
Chestnut. 


(d)  Equivalent  (effective)  tele- 
phone wire  or  power  wire  load 
(see  Fig.  4). 

(e)  Average  pole  spacing  (de- 
pends on  type  and  size  of  wire, 
usage,  loading  area,  and  class  of 
line  for  open  wire) . 

(/)  Length  of  poles  (deter- 
mined by  load  and  clearance  re- 
quirements). 

If  tables  are  not  available  for  determinrng  directly  the  class  of  poles  required  under  the 
above  known  or  assumed  conditions,  the  following  formulas  may  be  used  for  the  purpose. 

For  transverse  loading  (pole  acting  as  a  cantilever  beam) 


Lodge  pole  pine . 

Juniper 

Cypress 


3600 
5600 
7400 
6000 
6600 
4600 
5000 


c  = 


0.0002&4/£/ 


where  C  —  minimum  required  ground  line  circumference  of  new  pole  in  inches. 

Mw  —  resistant  moment  on  pole  at  ground  line,  due  to  wind  pressure  on  wires,  in 

ft-lb. 

-  PDLNS. 

P  SB  wind  pressure,  in  Ib/sq  ft. 

D  =  diameter  of  each  wire,  in  ft,  including  ice  coating,  if  any. 
L  »  distance,  in  ft,  between  center  of  the  load  and  the  ground  line  section. 
N  =  effective  number  of  wires  on  the  pole. 
S  =  1/2  sum  of  the  two  adjacent  spans,  in  ft. 
Mp  —  resistant  moment  on  pole  at  ground  line,  due  to  wind  pressure  on  pole,  in  ft-lb. 


where  h  =  height  of  pole  above  ground,  in  ft. 
Ct  —  circumference  of  pole  at  top,  in  in. 
Cg  =  circumference  of  pole  at  ground  line,  in  in. 
U  =  maximum  percentage  of  ultimate  fiber  stress  expressed  as  a  decimal  (determined 

from  the  line  classification,  Fig.  1,  or  as  required)  . 
/  ==  maximum  allowable  fiber  stress  of  the  pole  timber,  in  Ib/sq  in. 
For  vertical  loading  (usually  considered  only  for  anchor  guyed  poles  or  stubs,  where  the 
vertical  component  of  the  stress  in  the  guy  may  be  large),  the  pole  is  considered  to  be  a 
long  column,  and  by  Euler's  formula  for  such  a  column 

P.-MBg 

where  Pv  —  vertical  load  on  the  pole,  in  Ib. 

JLI  =  x2  for  average  conditions  of  a  guyed  pole  or  stub. 

E  —  modulus  of  elasticity  of  the  pole  timber. 

/  =  moment  of  inertia  of  the  critical  section  (see  note)  . 

I  —  length  of  the  column,  in  in.,  from  the  point  of  anchor  guy  attachment  to  the 
butt  of  the  pole  (for  poles  set  in  solid  bases,  such  as  rock  or  concrete,  the 
length  is  taken  from  the  point  of  anchor  guy  attachment  to  the  ground 
line  and  &  =  27T2). 

Note:  The  critical  section  in  flexure  for  the  average  pole  (of  conical  shape)  is  assumed 
to  be  at  a  distance  of  Vs  I  below  the  point  of  anchor  guy  attachment.  The  circumference 
of  this  section  should  be  computed  from  (1)  its  distance  above  a  point,  which  is  6  ft  from 
the  butt  of  the  pole,  (2)  the  specification  circumference  6  ft  from  the  butt,  and  (3)  the 
average  circumferential  taper  for  the  timber  under  consideration. 

In  general,  the  size  of  line  pole,  selected  for  the  ultimate  load,  is  determined  by  the 
transverse  stress,  to  which  the  pole  may  be  subjected.  There  may  be  cases,  however, 


10-54  TRANSMISSION  CIRCUITS 

where  the  vertical  load,  particularly  where  large  transformers  are  mounted  near  the  top 
of  the  pole,  becomes  a  substantial  factor  in  determining  the  pole  strength  required. 

In  such  cases,  both  transverse  and  vertical  stresses  and  their  resultant  stress  may  need 
to  be  determined,  although  the  vertical  and  combined  stresses  can  only  be  calculated 
approximately. 

A  wood  pole  is  considered  as  a  long  tapering  column,  which,  under  a  critical  concentric 
vertical  load,  fails  by  buckling.  The  maximum  load  that  the  pole  can  safely  carry  de- 
pends on  the  degree  of  freedom  of  the  pole  ends,  the  distribution  and  arrangement  of  the 
vertical  loading  on  the  pole,  the  support  given  the  pole  by  any  wire  or  cable  attachments, 
and  other  variable  factors. 

For  wood,  the  maximum  compressive  strength  is  considerably  less  than  the  modulus  of 
rupture  and  the  maximum  allowable  stress  for  the  combined  compression  and  bending 
stress  is  intermediate  between  these  two  values. 

Methods  have  been  developed  for  determining  approximately  the  combined  stress  on 
a  pole  at  the  ground  line,  due  to  vertical  (axial)  and  transverse  loads,  under  certain  as- 
sumptions, but,  because  of  their  complexity,  these  methods  are  not  discussed  in  this 
handbook. 

Where  it  is  desired  to  employ  such  methods,  reference  may  be  made  to  Provisional 
Report  Xo.  24  (Technical  Report  2G-2),  A  Study  of  Pole  Strength  in  Jointly  Used  Poles, 
of  the  Joint  Subcommittee  on  Development  and  Research  of  the  E.E.I,  and  B.T.S.  of 
August  22,  1938. 

The  minimum  required  ground-line  circumference  of  the  pole  C  having  been  determined, 
the  class  of  pole  actually  selected  from  the  ASA  Specification  Tables  should  have  a  ground- 
line  circumference  at  least  equal  to  the  minimum  required  ground-line  circumference  C 
plus  an  amount  of  wood  which,  based  on  average  decay  rates  for  the  timber  and  location 
involved,  will  provide  the  desired  service  life  in  accordance  with  the  formula 

C,  -  C  +  y(L  -  T) 
where  Cg  =  minimuni  ground-line  circumference  of  the  new  pole,  in  in.,  to  provide  desired 

service  life. 
C  —  minimum  required  ground-line  circumference  of  new  pole,   determined  as 

above. 

y  —  average  rate  of  decay  of  untreated  timber  in  equivalent  inches  of  circumference 
per  year.     (For  the  cedars  and  chestnut,  this  decay  is  about  0.45  in.  for 
sap  wood  and  0.3  in.  for  heartwood  per  year.    For  southern  pine  the  physical 
life  is  about  equal  to  the  effective  period  of  treatment.) 
L  —  desired  physical  life  in  years. 
T  =  expected  life  of  preservative  treatment  in  years  (probably  roughly  20  years 

for  initial  treatment  of  butt-treated  poles). 

The  ASA  (American  Standards  Association)  Specification  Tables  of  Pole  Dimensions, 
referred  to  above,  are  readily  available  and  are  based  primarily  on: 

1.  Fiber  strengths  of  various  timbers  used. 

2.  Ten  classes  of  poles  (1  to  10)  with  Tm'mrrmm  circumferences  for  6  ft  from  the  butt 
specified  for  classes  1  to  7,  and  minimum  circumferences  at  the  top  of  the  pole  specified 
for  all  classes. 

3.  Breaking  loads  2  ft  from  the  top  for  the  first  seven  classes  in  approximate  geometric 
progression  as  follows: 

Class  Breaking  Load,  in  Ib 

1  4500 

2  3700 

3  3000 

4  2400 

5  1900 

6  1500 

7  1200 

4.  All  new  poles  of  the  same  class  and  length  (classes  1  to  7  only)  to  have  about  equal 
resistant  moments  at  the  ground  line. 

5.  All  new  poles  of  the  same  class  (classes  1  to  7  only)  to  be  of  such  size  as  to  have  about 
the  same  breaking  load,  with  the  load  applied  2  ft  from  the  pole  top  and  assuming  that 
the  break  would  occur  at  the  ground  line. 

Treatment  of  poles  to  prolong  their  physical  life  is  standard  practice.  The  preservatives 
used  may  be  creosote,  greensalt,  or  other  chemicals  toxic  to  wood-destroying  fungi  and 
wood-boring  insects.  The  creosote  treatment  may  be  applied  the  full  length  of  the  pole, 
as  in  southern  pine,  under  a  pressure  and  vacuum  with  a  net  retention  normally  of  6  to  8 
Ib  of  creosote  per  cubic  foot  of  wood,  and  a  penetration  of  not  less  than  2  1/2  in.  or  85  per 
cent  of  the  sapwood.  Greensalt  is  also  applied  to  poles  by  the  pressure  method,  and  it 
has  some  advantages  over  the  creosote  treatment.  Butt-treated  poles,  such  as  the  cedars, 


TRANSMISSION-LINE  CONSTRUCTION 


10-55 


are  usually  processed  in  open  tanks  of  hot  and  then  cold  creosote,  the  ground-line  section 
being  incised  to  assist  penetration,  which  averages  about  0.4  in.  or  more. 

The  spacing  of  poles  depends  largely  upon  the  type  of  the  wire  or  cable  load,  location, 
transposition  scheme,  and  exposure  to  storms.  For  important  backbone  toll  routes  the 
spacing  is  generally  about  130  ft  for  open  wire  and  from  150  to  300  ft  for  toll  cable. 

The  guying  of  pole  lines  is  necessary  at  points  of  above  average  stress,  such  as  at  corners 
having  substantial  pulls,  dead-end  poles,  and  poles  carrying  unusual  loads.  Guying  is 
also  applied  on  toll  lines  at  periodic  intervals  along  the  line  to  assist  the  pole-line  structure 
in  withstanding  storms.  The  ratio  of  lead  to  height  of  guys  should  be  about  1.0  to  1.25, 
the  guy  stress  then  being  about  1.4  to  1.28  times  the  horizontal  stress. 

Cross-arms  are  designed  to  carry  from  2  to  10  or  12  wires,  as  may  be  required  for  any 
particular  case,  and  may  be  fitted  with  locust  or  steel  pins,  usually  spaced  from  6  to  12 
in.  for  telephone  and  10  and  11  1/4  in.  for  telegraph  wires  (16  to  30  in.  for  pole  pairs).  The 
cross-arms  are  usually  spaced  24  in.  apart  vertically  on  the  pole  and  vary  in  number  per 
pole  from  1  to  6  or  more. 

OPEN  WIRE,  supported  on  insulator-equipped  cross-arms  or  brackets,  which  are  in 
turn  mounted  on  poles  or  fixtures,  has  been  employed  since  the  invention  of  the  telegraph 
and  telephone  to  connect  individual  instruments  to  wire  centers  or  offices,  and  one  office 
to  another.  However,  wire  conductors  enclosed  in  lead  sheaths  (cables)  have  practically 
superseded  open  wire  in  built-up  communities  and  cities  and  in  large  measure,  for  tele- 
phone communication,  have  replaced  or  supplemented  open-wire  lines  between  the  prin- 
cipal cities. 

The  types  of  open  wire  employed  for  telephone  communication  circuits,  as  discussed  in 
Section  17,  consist  principally  of  104,  128,  and  165  hard-drawn  copper  and  some  104  and 
128  copper  steel  (40  per  cent  cond.)  for  toll  circuits,  080  copper  steel  (40  per  cent  cond.), 
and  080  and  109  high-tensile  steel  for  exchange  circuits  in  outlying  areas  (all  given  in  mil 
diameter).  High-tensile  steel  wire,  because  of  the  pole  economies  realized,  its  greater 
strength,  and  its  equally  good  service  performance,  is  being  employed  for  new  construction 
generally  in  place  of  the  various  grades  of  mild  steel  and  iron  wire. 

The  types  of  open  wire  employed  for  telegraph  communication  circuits  consist  mainly 
of  114  hard-drawn  copper,  162  copper  steel  (40  per  cent  cond.),  and  some  165  iron  for 
important  facilities. 

Drop  wires  of  various  types  are  used  for  both  services. 

Table  2  shows  some  of  the  important  physical  properties  and  the  electrical  resistance 
of  wire,  classed  as  open  wire,  for  both  telephone  and  telegraph  circuits.  Other  electrical 
characteristics  of  various  types  of  wire  are  discussed  in  Section  17. 

Table  2 


Type  of  Wire 

Wire 
Number 
and 
Gage 

Nominal 
Wire 
Diameter, 
in  mils 

Average 
Weight 
per  Wire, 
in  Ib/mi 

Minimum 
Breaking 
Strength 
per  Wire, 
inlb 

Average 
Resistance  per 
Loop  Mile, 
in  ohms 
at  68°  F 

Telephone 
Hard-drawn  copper 

14-NBS 

80 

102 

330 

17  50 

Hard-drawn  copper  

12-NBS 

104 

173 

550 

10.  15 

Hard-drawn  copper            .    . 

10-NBS 

128 

262 

819 

6  74 

Hard-drawn  copper  

8-BWG 

165 

435 

1325 

4.11 

Copper  steel  (40%) 

14-NBS 

80 

96 

770 

42  8 

Copper  steel  (40%)  

1  2-NBS 

104 

159 

1177 

25.0 

Copper  steel  (40%)  

10-NBS 

128 

240 

1647 

16.7 

High-tensile   steel    (HTL-85) 
(0.8  oz  zinc  coating)  
(HTL-135)    

14-BWG 
14-BWG 

83 
83 

99 
99 

460 
703 

117.2 
130.0 

(HTL-85)  

1  2-BWG 

109 

170 

793 

68.2 

(HTL-135) 

1  2-BWG 

109 

170 

1213 

76.5 

(HTL-85)   . 

10-BWG 

134 

258 

1199 

45.0 

Bronze  TP  drop  

18-\WG 

40 

159 

340 

259.0 

Bronze  TR  drop   .... 

18-4.WG 

40 

232 

340 

259.0 

Bronze  NP  drop 

18-4.WG 

40 

227 

340 

259.0 

Hard-drawn  copper  HC  drop. 

Telegraph 
Hard-drawn  copper 

14-AWG 
9-AWG 

64 
114 

316 
208 

380 
644 

26.4 
4.3 

Copper  steel  (40%)           

6-AWG 

162 

384 

2430 

5.3 

Iron  

8-BWG 

165 

378 

1090 

13.3 

Copper  (tw.  pr.)  

1  6-AWG 

51 

208  (pr.) 

500  (pr.) 

104.0  (pr.) 

Steel  drop  (sgl.)  

1  6-AWG 

51 

104  (sgl.) 

250  (sgl.) 

52.0  (sgl.) 

10-56 


TRANSMISSION  CIRCUITS 


CABLE,  consisting  of  insulated,  annealed  copper  conductors,  enclosed  in  a  cylindrical 
lead  sheath,  is  employed  in  both  toll  and  exchange  telephone  and  in  telegraph  plant,  where 
the  number  of  circuits  required  along  a  given  route,  plant  economies,  or  interfering  condi- 
tions preclude  the  use  of  open  wire.  Cable  is  used  generally  in  toll  plant  in  urban  areas 
and,  supplementing  open  wire,  between  principal  traffic  centers  throughout  the  country, 
either  as  toll  entrance  facilities  for  open-wire  lines  or  as  toll  cable  facilities  directly  connect- 
ing large  switching  centers. 

Telephone  and  telegraph  cable  is  manufactured  in  various  sizes  and  gages,  of  which 
representative  types  and  associated  data  are  given  in  Table  3.  For  both  types  of  service, 

Table  3.    Representative  Cables — Mechanical  Characteristics 


Type 

(3) 
No.  of 
Pairs 

(1)  (2) 
Gage 
of  Con- 
ductor 

Conductor 
Insulation 

Sheath 
Thick- 
ness, 
in. 

Outside 
Diameter, 
in. 

Weight 
per 
Foot, 
Ib 

Type  of 
Core 

Exchange  

6 

19 

Paper  tape 

0.063 

0.42 

0.41 

Layer 

Exchange 

455 

19 

Paper  tape 

0.115 

2.61 

8.48 

Layer 

Exchange  -  , 

11 

22 

Wood  pulp 

0.063 

0.42 

0.40 

Layer 

Exch&ngj? 

909 

22 

Wood  pulp 

0.115 

2.61 

8.46 

Multiple-^unit 

Exchange  -  «  -    .  r  - 

11 

24 

Wood  pulp 

0.061 

0.36 

0,31 

Layer 

TSxrhft,isgp 

1515 

24 

Wood  pulp 

0.115 

2.61 

8.64 

Multiple-unit 

Exch&ngc   .         ... 

11 

26 

Wood  pulp 

0.060 

0.32 

0.27 

Layer 

ExrhiAnge       ,    .  -  -  - 

2121 

26 

Wood  pulp 

0.115 

2.61 

8.15 

Multiple-unit 

Toll  entrance 

(4) 
12  quads 

19 

Paper  tape 

0.082 

0.85 

0.79 

Layer 

Toll  entrance 

i    3  quads 
\  27  ouads 

13 
16 

Paper  tape  > 
Paper  tape  > 

0.  118 

2.35 

6.41 

Layer 

160  quads 

19 

Paper  tape  J 

Toll  (full  size)  

154  quads 

19 

Paper  tape 

O.T23 

2.59 

7.60 

Layer 

Notes;  (I)  All  conductors  are  annealed  copper. 

(2)  Some  28  gage  exchange  cable  was  made  during  World  War  II  to  conserve  copper  in  sizes 
11  to  303  pairs. 

(3)  The  smallest  and  largest  sizes  of  exchange  cable  are  shown,  as  used  in  the  Bell  System. 

(4)  Toll  entrance  and  toll  cable  may  have  optional  groups  of  non-quadded  exchange  pairs. 
The  cable  data  given  assume  no  shielding  or  sheath  protection  or  exchange  conductors. 

(5)  Insulation  resistance  required  to  exceed  500  megohms  per  mile. 

(6)  Some  13  gage  is  used  in  both  telephone  and  telegraph  service. 

paper  or  wood-pulp  insulated  conductors  are  grouped  together  to  form  a  core,  either  in 
layers  or  51  and  101  pair  units,  various  colors  being  used  in  the  insulation  and  binding 
strings  to  permit  readily  distinguishing  between  different  layers,  units,  quads,  or  pairs,  for 
installation  or  maintenance  purposes.  Figure  5  shows  the  method  of  core  construction 
for  both  layer  and  multiple-unit  type  cores  of  telephone  exchange  cable  (24  gage-type 
DSM). 

Exchange  conductors  are  generally  associated  in  pairs  (2  conductors  twisted  together), 
although  some  exchange  trunk  cable  is  quadded  (2  pairs  twisted  together  as  a  4-conductor 
unit  group,  called  a  quad).  Toll  conductors  are  usually  quadded  for  phantom  circuit 
operation,  although  toll  cables  frequently  contain  some  non-quadded  pairs  for  program 
or  other  toll  services.  In  addition,  complements  of  exchange  pairs  are  frequently  included 
in  the  toll  cable.  The  lead  sheath  enclosing  the  paper-wrapped  core  of  insulated  conductors 
normally  contains  about  1  per  cent  of  antimony  to  strengthen  the  sheath,  the  thickness  of 
which,  varies  with  core  diameter  and  type  of  cable  from  about  0.06  to  0.125  in. 

Cross-talk  must  be  carefully  considered  in  cable  design,  particularly  for  toll  cables.  In 
non-quadded  cables  the  conductors  are  twisted  together  in  pairs.  The  twists  in  adjacent 
pairs  within  a  layer  and  in  adjacent  layers  are  of  different  lengths  to  provide  the  necessary 
capacitance  balance  between  pairs.  In  quadded  cables  the  conductors  are  twisted  together 
in  pairs  and  the  pairs  are  twisted  together  with  different  lengths  of  twist  to  form  quads 
of  as  many  as  9  types,  each  of  which  has  a  different  length  of  twist.  Exchange  pairs  (non- 
quadded)  are  usually  random  spliced  within  their  color  groups  (spliced  without  testing 
t®  <ie*ermiBe  thB  pair  numbers),  so  that  any  given  pair  in  the  cable  is  adjacent  to  any  other 


TKANSMISSION-LINE  CONSTRUCTION 


10-57 


pair  in  as  few  cable  sections  as  practicable.  Toll  pairs,  because  of  their  greater  importance, 
wider  range  of  operating  energy  levels  and  frequency  assignments,  are  carefully  spliced 
where  sections  of  cable  join.  The  splicing  is  carried  out  according  to  a  plan  which  provides 
for  limiting  the  cross-talk  between  circuits. 

Segregation,  to  reduce  couplings  at  carrier  frequencies  between  oppositely  transmitting 
pairs  or  quads  used  for  cable  carrier,  is  usually  accomplished  by  assigning  oppositely  bound 
groups  to  separate  cables.  Segregation  of  pairs  used  for  open-wire  carrier  is  obtained  by 
using  alternate  layers  or  by  metallic  layer  or  unit  quad  shields  in  the  same  cable,  as  may 
be  required  to  prevent  excessive  cross-induction  between  them. 

Sheath  protection  is  provided  in  various  degrees,  as  may  be  required,  by  using:  (1)  layers 
of  Sisalkraft  paper  and  jute  covering,  where  soil  corrosion  may  occur;  (2)  layers  of  thermo- 
plastic compound  covered  by  longi- 
tudinal copper  tape,  flooded  with 
asphalt  compounds,  for  a  lightning 
shield;  (3)  layers  of  Sisalkraft  paper, 
two  steel  tapes,  and  jute,  for  gopher- 
infested  areas;  (4)  layers  of  Sisalkraft 
paper  and  a  layer  of  rubber  or  as- 
phalt-back fabric  tape  for  corrosion 
protection  in  conduit;  (5)  a  thermo- 
plastic compound  layer  and  outer 
covering  of  impregnated  fabric  tape 
for  corrosion  protection  when  buried 
near  pipe  lines;  (6)  two  helical  wrap- 
pings of  tape  armor  and  a  cushion  of 
jute  to  protect  against  ring  cutting, 
stone  bruises,  or  to  provide  shielding 
for  low-frequency  induction  for  aerial 
cables;  and  (7)  single  and  double 
armor  wires  with  a  jute  cushion  for  _/  eoepwjRs 
submarine  cables.  Other  special 
protection  may  be  provided  as  re- 
quired. 

Spiral-four  disk-insulated  cable  is 
used,  under  certain  conditions,  to 
provide  entrance  facilities  for  open- 
wire  carrier  systems  operating  at  a 
maximum  frequency  of  about  140 
fcc.  The  individual  units  contain  4 
copper  conductors  of  16  gage,  insu- 
lated from  each  other  by  composition 

disks  having  uniformly  spaced  peri-  ££  l&fSSSSSgcSSS  <J 
pheral  notches  which  hold  the  wires 
spaced  at  the  corners  of  a  square,  each  oppositely  positioned  pair  of  wires  forming  a  pair 
of  the  quad.  Shielding  is  provided  over  a  spiral-4  unit  consisting  of  a  copper  tape  and  two 
steel  tapes.  A  toll  entrance  cable  may  contain  up  to  7  such  units,  each  with  its  own  metal 
sheath,  or  a  combination  of  units  and  standard  paper-insulated  quads  and  pairs  which 
provide  other  communication  facilities. 

Coaxial  cable,  of  latest  design,  for  use  with  the  carrier  and  other  high-frequency  systems, 
consists  of  up  to  8  units.  Each  unit  is  composed  of  a  100.4-mil  conductor  (inner  conductor) 
positioned  in  the  center  of  a  single  12-mil  copper  tape  (outer  conductor)  with  longitudinal 
seam  to  form  a  0.375-in.  (inner  diameter)  tube,  over  which  are  lightly  wrapped  two  6-mil 
steel  tapes.  The  outer  conductor  is  supported  by  polyethylene  disks,  0.085  in.  thick, 
spaced  about  1  in.  apart  along  the  inner  conductor.  The  outer  and  inner  conductors  form  a 
metallic  circuit.  With  the  8  units,  other  standard  paper-insulated  quads  of  conductors 
(up  to  about  78  quads  of  19  gage  or  equivalent)  may  be  included  in  the  same  overall  lead 
sheath.  Further  details  regarding  this  cable  and  its  characteristics  are  given  in  Section  17. 

Both  exchange  and  toll  cables  are  placed  aerially  and  underground,  depending  upon  the 
location  and  other  factors  affecting  any  given  cable. 

When  cables  are  placed  aerially  on  poles  or  towers,  the  required  strength  of  the  support- 
ing structures  must  be  carefully  determined  on  the  basis  of  loading  area,  exposure  to 
weather  conditions,  load  carried,  importance  of  the  cable  to  service,  and  other  factors  such 
as  apply  in  open-wire  construction.  When  cables  are  placed  underground,  the  main  factors 
to  consider  are  the  type  of  underground  housing,  if  any,  to  employ,  as  well  as  the  possibili- 
ties of  damage  from  corrosion  or  other  external  sources,  and  the  proper  routing.  Under- 


KEY: 

W-C  =  WHITE -GREEN 
W-B  =  WHITE- BLUE 

W-RsWHITE-RED 
(?)  =BLU6-RED  TRACER  ftMR 

NOTE. 

NUMBERS  ARE  TOTAL  PAIRS  IN  CROUP 
INCLUDING  ONE  TRACER  BMR  WHERE 
INDICATED 

ALL  UNITS  JN  THE  SAME  LAYER  HAVE 
UKE  COLORED  BINDING  STRINGS 

Diagrams  Showing  Typical  Layer  and  Multiple- 
"     '  ~  "     ~          (Courtesy  Bell  System) 




e BWRS 


10-58  TRANSMISSION  CIRCUITS 

ground  cables  may  be  placed  in  vitrified  clay  conduit,  fiber,  or  other  types  of  ducts  or 
buried  directly  in  a  trench  in  the  ground  or  plowed  into  the  ground.  In  any  event,  suitable 
protection  from  underground  hazards  must  be  provided. 

Loading  coils  are  provided  in  cable  circuits  (aerial  or  underground)  wherever  required 
for  transmission  reasons  (see  Section  17). 

Loading  coils  are  generally  assembled  in  groups  in  steel  cases  for  either  aerial  or  under- 
ground installations,  and  some  designs  are  suitable  for  office  relay  rack  mounting.  Lead 
sleeve  cases  are  used  for  loading  small  complements  of  toll  and  exchange  conductors.  Also 
single  coils  (in  a  small  metal  case)  are  designed  to  be  connected  and  enclosed  in  cable 
splices. 

UNDERGROUND  STRUCTURES,  consisting  of  conduit,  manholes,  vaults,  and  other 
construction,  are  designed  to  provide  suitable  housings  for  underground  cable  plant.  These 
structures  are  a  necessity  in  cities  and  to  some  extent  in  the  smaller  communities  (assuming 
that  aerial  cable  plant  is  not  practicable  or  economical  in  a  given  case) ,  since  the  under- 
ground cable  plant  must  be  readily  accessible  to  permit  additions,  changes,  removals,  and 
repairs.  Some  underground  exchange  cable  has  been  buried  (placed  directly  in  the  ground) 
in  built-up  locations,  but  this  is  not  the  general  practice.  Toll  cables,  suitably  protected, 
are  frequently  plowed  into  the  ground  or  are  buried  in  trenches.  This  type  of  construction 
has  been  employed  for  long  distances  over  transcontinental  and  other  important  toll  cable 
routes. 

Underground  conduit  may  be  of  vitrified  clay,  fiber  composition,  creosoted  wood,  or 
iron  pipes.  Clay  conduit  is  most  generally  employed  for  main  routes  because  of  its  rela- 
tively low  cost  and  satisfactory  performance  when  buried  and  properly  protected.  It  is 
manufactured  in  single  or  2-,  3-,  4-,  6-,  8-,  and  9-duct  multiple  units,  and  is  usually  laid 
in  one  or  more  units  in  a  compact  arrangement.  The  conduit  is  placed  in  a  trench,  which 
may  vary  in  depth  to  avoid  other  underground  structures  or  hazards  (sharp  changes  in 
direction  are  avoided),  on  a  solid  base  with  or  without  concrete.  It  may  be  encased  com- 
pletely in  concrete  or  only  at  the  top  or  bottom,  or  plank  may  be  used,  in  the  judgment  of 
the  engineer  as  to  the  type  of  construction  needed. 

Conduit  may  be  placed  along  a  street  under  paving  or  the  sidewalk,  or  between  the  curb 
and  sidewalk,  or  in  parking  in  the  center  of  the  street,  as  conditions  permit.  Costs  and  a 
satisfactory  permanent  location,  least  likely  to  be  disturbed,  are  primary  considerations 
in  placing  conduit  over  a  selected  route.  Subsidiary  ducts  of  wood  and  of  vitrified  clay, 
sewer  or  iron  pipe  are  commonly  used  from  manholes  to  underground  cable  poles  or  build- 
ings. 

Manholes  are  required  at  junction  points  of  conduit  runs  and  other  locations  where  it 
is  desirable  that  the  underground  cable  plant  be  made  accessible,  and  to  provide  for 
practicable  cable  section  lengths.  Manholes  are  preferably  located  at  one  side  or  the  other 
of  street  intersections  to  avoid  interfering  with  traffic,  when  working  in  them.  Manholes 
vary  in  size  from  the  small  service  boxes,  placed  mainly  for  the  purpose  of  pulling  cable, 
to  the  standard  2-,  3-,  and  4-way  and  center  rack  types,  having  dimensions  ranging  from 
3  ft  6  in.  width  by  6  ft  length  by  5  ft  6  in.  headroom  to  8  ft  width  by  9  ft  length  by  about 
6  ft  or  more  headroom.  In  addition,  special  type  manholes  and  cable  vaults  of  various 
sizes  are  required  in  many  cases  to  accommodate  concentrated  conduit  entrances,  as  at 
central  offices  and  loading  coil  installations.  Concrete  construction  is  usually  employed, 
with  or  without  reinforcement.  Manhole  frames  and  covers  of  cast  iron,  placed  in  the 
manhole  roof  to  provide  a  suitable  entrance  of  27-in.  diameter  or  more  to  the  manhole, 
are  designed  to  support  surface  traffic  safely.  Drainage  may  or  may  not  be  provided  in 
manholes,  as  required. 

The  underground  cables  are  supported  on  galvanized-metal  racks,  mounted  vertically 
along  the  sides  or  through  the  center  of  the  manhole,  and  are  separated  by  a  few  inches  to 
permit  splices  to  be  made  and  opened  as  required. 

26.  ELECTRICAL  PROTECTION  OF  TRANSMISSION  LINES 

Lightning  and  power  circuits  are  two  unlike  sources  of  electrical  power,  which,  under 
certain  conditions  of  exposure  or  contact  encountered  in  practice,  occasionally  cause  dam- 
age to  communication  plant. 

Communication  lines  and  equipment  are  necessarily  grouped  closely  together  because 
of  space  limitations  and  other  economic  considerations.  These  facilities  will  carry  their 
normal  operating  voltages  and  currents  with  ample  margins  of  safety,  but  the  insulation 
employed  on  cable  conductors,  between  conductors  and  the  cable  sheath,  and  in  various 
equipments  is  not  sufficient,  in  general,  to  withstand  lightning  or  power  circuit  potentials 
witliout  protection. 


\ 


ELECTRICAL  PROTECTION  OF  TRANSMISSION  LINES      10-59 

EXCHANGE  CABLE  PROTECTION  against  lightning  for  any  given  exchange  cable 
plant  usually  requires  a  study  of  the  plant  layout  and  exposure  conditions  involved.  The 
aerial  cable  sheath  is  usually  grounded  through  the  underground  cable  plant  or  at  the  cen- 
tral office  and  at  various  other  locations  at  irregular  intervals,  such  as  at  underground  dips 
or  private  cable  entrances. 

The  working  cable  conductors  are  grounded  by  operation  of  the  station  protection  and 
through  the  loops  connecting  to  them.  Cable  conductors  may  serve  subscriber  stations 
at  one  or  more  intermediate  points  along  the  conductors  but  lie  idle  in  other  sections  of 
the  cable  beyond  these  points.  Other  conductors  may  lie  idle  throughout  their  length, 
and  groups  of  conductors  will  terminate  at  various 
points  where  cable  sizes  change.  All  these  conditions 
have  a  direct  bearing  on  potentials  in  exchange  cables 
due  to  lightning. 

Lightning  currents  may,  in  general,  be  impressed  on 
exchange  cables  by  (1)  inductive  coupling  between  the 
cables  (or  between  conductors  connected  to  them)  and 
the  lightning  path,  (2)  direct  stroke,  or  (3)  metallic  con- 
nection with  entering  open-wire  or  drop  loops,  carrying 
lightning  potentials,  as  a  result  of  a  direct  stroke  or 
other  causes.  Lightning  currents  from  metallic  connec- 
tions (by  conduction)  are  usually  the  principal  con- 
sideration in  the  design  of  exchange  cable  protection. 
Lightning  current  may  appear  on  the  open  wire,  con- 
nected to  the  cable,  from  such  sources  as  direct  strokes 
to  the  line,  conduction  from  service  drops  or  guys,  or 
from  arcs  from  power-system  ground  wires.  Figure  6 
shows  the  results  of  crest  current  measurements  at  ex- 
change cable-open  wire  junctions. 

Although  lightning  currents  reach  cables  over  both  connected  drop  and  open-wire  loops, 
it  appears  that  the  currents  on  the  drop  loops  will  probably  be  greater  than  those  on  the 
open  wire.  However,  cable  plant  with  drop  loops  is  generally  located  in  built-up  areas 
having  low-resistance  public  water  systems  and  various  types  of  shielding,  whereas  open- 
wire  plant  extends  mostly  into  rural  areas  where  direct  lightning  strokes  are  more  likely 
to  occur  and  where  grounding  conditions  are  not  as  favorable  as  in  built-up  areas.  Thus, 
cable  damage  from  lightning  currents  is  more  likely  to  result  from  open-wire  than  from  drop 
loops.  In  some  locations  where  cables  extend  into  rural  districts  and  are  subject  to  the 
same  lightning  exposure  as  the  open  wire  the  current  received  by  the  cable  from  the  drop 

loops  may  be  of  greater  importance  than  that  re- 
ceived  from  the  open-wire  loops  connected  to  these 
cables. 

Lightning  currents,  reaching  exchange  cable  from 
any  source,  produce  potential  differences  between 
conductors  and  between  the  sheath  and  conductors 
which,  if  protection  is  not  provided,  may  seriously 
damage  the  sheath  and  the  enclosed  conductors. 

The  potential  differences  within  the  cable,  result- 
ing from  current  on  the  conductors,  depend  largely 
on  the  current  wave  form.  For  steep  wave-front 
surges,  the  relatively  high  self  and  mutual  surge 
impedances  are  important  in  determining  the  poten- 
tial differences;  for  slower  surges  or  short  cables 
(about  1  to  2  miles)  these  potential  differences 
depend  more  upon  the  conductor  resistances.  Fre- 
quently, the  slower  surges  produce  the  lower  poten- 
tial differences.  Figure  7  illustrates  the  voltage  dis- 


O  20  4O  6O  8O  JOO 

PER  CENT  OF  CUfWEMT    EXC££DiN« 

ORDi  NATES 

FIG.  6.     Crest  Current  Distribution 
for  Lightning  Currents  at  Exchange 
Cable — Open  Wire  Junctions  (Cour- 
tesy Bell  System) 


•US.  CABLE  ••* AERIAL   CABLE 


ET 


wx/wwiw/a     SHEATH   _ 

i 
CONDUCTOR   GROUPS: 

1.  CONNECTED  TO  WJRE 

2.  SPARE 
3.WOftKJNft  AT  X 

\ 


FIG.     7. 


DISTANCE    ALONG   CABLE 

Voltage     Distribution 


^  _^.     _w        i_j_ along 

Conductors  andlSheath  for  Current  De- 
livered from  Open  Wire — No  Protector 
Blocks   on   Conductors    (Courtesy   Bell 
System) 

tribution  along  conductors  and  sheath  for  current  delivered  from  an  open-wire  loop  with 
no  protector  blocks  at  the  junction  of  the  open  wire  and  cable.  The  aerial  cable  is  not 
grounded,  except  through  the  underground  section  and  the  office  ground,  and  contains  a 
group  of  conductors  (1)  connected  to  open-wire  loops,  (2)  spare,  and  (3)  working  at  an, 
intermediate  point,  x,  along  the  cable. 

Potential  differences  in  cables,  unless  several  times  the  breakdown  voltage  of  tne  cable 
insulation,  do  not,  in  general,  cause  permanent  damage,  although  the  insulation  may  be 
punctured  in  numerous  places.  Remedial  measures  are,  therefore,  designed  to  reduce  the 
applied  voltages  between  conductors  and  between  conductors  and  the  sheath  and  to  limit 
or  distribute  the  current  so  that  permanent  failures  are  unlikely  even  though  punctures 


10-60 


TRANSMISSION  CIRCUITS 


do  occur.  Such  measures  may  consist  of  (1)  connecting  protector  blocks  between  conduc- 
tors and  sheath  where  potential  differences  are  most  likely  to  be  high  and  occur  frequently, 
(2)  providing  means  of  diverting  from  the  cable  a  substantial  part  of  the  current  which 
might  reach  the  cable  over  open-wire  or  drop  loops  serving  installations  exposed  to  lightning 
strokes,  such  as  radio  and  fire  towers,  and  (3)  placing  conductors  parallel  with  the  sheath, 
thus  increasing  the  conductivity  and  reducing  the  IR  drop  along  the  sheath. 

Protector  Blocks.  One  type  of  protector  block,  commonly  employed  between  cable 
conductors  and  the  enclosing  sheath  at  the  junction  of  open-wire  or  drop  loops  and  cables, 
consists  of  a  porcelain  block  with  a  small  carbon  block  insert  held  in  place  by  a  quick- 
melting  glass  cement.  The  porcelain  block  is  held  against  a  solid  carbon  block  (which 
rests  against  a  grounded  metal  plate)  by  a  metal  spring  bearing  against  the  small  carbon 
insert.  The  insert  is  positioned  to  provide  an  air  gap  between  it  and  the  grounded  carbon 
block  of  about  0.006  in.  One  set  of  these  blocks  is  provided  for  each  of  the  two  conductors 
of  the  cable  pair.  When  the  current  in  the  air  gap  is  sufficient  to  melt  the  cement,  the 
metal  spring,  which  is  connected  to  a  cable  conductor,  forces  the  carbon  block  insert 
against  the  ground  block,  thus  grounding  the  loop  wire  and  cable  conductor  to  which  the 
loop  wire  is  connected. 

These  protector  blocks  break  down  when  steep  wave  front  voltages  of  1000  to  1500  volts 
are  impressed,  this  voltage  range  being  generally  somewhat  below  the  steep  wave  front 
voltage  necessary  to  puncture  exchange  cable  insulation.  Protection  is  thus  provided  for 
the  cable  conductors,  having  protector  blocks,  against  dielectric  failure  near  to  the  point 
where  these  blocks  are  connected.  However,  damaging  potential  differences  may  occur 
between  the  protected  conductors  and  other  conductors  at  the  point  of  protection  and  be- 
tween conductors  and  sheath  at  other  points.  Lower-breakdown  protector  blocks  for 
cable  protection  do  not  appear  to  offer  an  important  improvement  in  protection  and 
generally  react  unfavorably  from  the  standpoint  of  maintenance  and  service. 

Protector  blocks  applied  to  working  conductors  at  the  junction  of  open-wire  and  aerial 
cables  (not  grounded  at  intermediate  points)  reduce  potential  differences  between  such 

conductors  and  the  sheath  and  other  conductors,  as 
shown  by  a  comparison  of  Figs.  7  and  8.  The  curves 
in  these  figures  are  not  drawn  to  scale,  and  actually 
the  reductions  obtained  are  much  larger  than  can  be 
indicated  conveniently.  With  working  conductors 
thus  protected  (Fig.  8),  the  maximum  potential  differ- 
ence within  the  cable  usually  occurs  at  the  open-wire 
junction  between  the  sheath  and  spare  pairs,  and  the 
next  highest  potential  difference  appears  at  the  junc- 
tion of  the  aerial  and  underground  cable  between  the 
sheath  and  protected  conductors. 

When  protector  blocks  are  applied  to  all  conductors 
at  the  junction  of  open-wire  and  aerial  cables,  the 
potential  differences  between  conductors  are  elixni- 
Fro.    8.    Voltage    Distribution    along  nated  throughout  the  cable  (assuming  an  ideal  con- 
SSS^S^S?   Wir^ototo  tit™  of  no  Sounds  on  the  conductors  through  station 
Block    on    Conductors    Connected    to   blocks  at  other  points) .    The  only  potential  difference 
Wire  (Courtesy  Bell  System)  existing  in  the  cable  under  this  condition  is  between 

the  sheath  and  all  conductors,  being  a  maximum  at 

the  junction  of  the  aerial  and  underground  cable  sections  and  about  equal  to  the  IR  drop 
on  the  conductors  through  the  underground  section. 

Protector  blocks  (usually  with  10-mil  air  gaps)  for  all  wires  may  also  be  placed  along 
the  open-wire  line,  as  required,  to  divert  current  from  the  cable.  This  results  in  increased 
effectiveness  for  a  given  ground  impedance,  due  to  increasing  the  impedance  between  the 
ground  point  and  the  underground  cable,  over  what  it  would  be  for  a  single  sheath  ground 
at  the  open-wire  junction,  and  it  avoids  the  adverse  effects  of  a  single  ground  on  the  sheath. 
Experience  indicates  that  improvements  usually  result  when  the  open-wire  protection  is 
placed  within  about  700  to  3000  ft  of  the  cable  terminal  and  the  ground  resistance  at  the 
protection  does  not  exceed  about  30  ohms. 

Grotmds  may  consist  of  several  hundred  feet  of  buried  wire  in  high-resistivity  areas,  or 
rods  may  be  used  effectively  in  low-resistivity  areas.  Grounds  of  limited  extent  have 
impedances  about  equal  to  their  d-c  resistances,  but  for  lengths  of  wire  or  pipe  greater 
than  about  200  ft  the  d-c  resistance  decreases  with  length  faster  than  the  impedance. 
Figure  9  shows,  for  homogeneous  earth,  the  variation  of  resistance  with  length  of  buried 
wire,  the  earth  resistivity  being  100  meter  ohms.  Large  variations  may  be  expected  in 
tlieae  values  where  the  soil  has  a  highly  variable  resistivity.  For  grounds  of  equal  d-c 
resistances,  the  ground  of  least  extent  is  considered  best  for  grounding  purposes;  although, 


CONDUCTOR  SROOPS: 
J.  CONNECTED  TO  WWE 


DISTANCE   ALONC   CABLE 


ELECTRICAL  PROTECTION  OF  TRANSMISSION  LINES      10-61 


*PG*  WO  METER-OHMS  EARTH  RESISTIVITY 


25      50     75      tOO     300  50O  TOO   900 

LENGTH   OF  WIRE  IN   FEET 

FIG.    9.     Approximate    Resistance    to 

Ground  of  a  Buried  Wire   (Courtesy 

Bell  System) 


practically,  the  neutral  of  a  well-grounded  common  neutral  power  system  or  water  piping 
system,  if  available,  will  probably  provide  a  better  ground  than  a  made  ground. 

Shielding  conductors  placed  on  the  pole  line  along  with  the  cable  or  buried  below  it 
will  reduce  conductor-to-conductor  and  sheath-to-conductor  potentials  by  about  50  per 
cent  or  more.  The  effectiveness  of  shielding  depends  upon  diverting  part  of  the  current 
from  the  sheath.  This  method  of  current  diversion  has  been  limited  because  more  prac- 
ticable methods  are  available. 

Increasing  sheath  conductivity  by  placing  bare  conductors  along  the  sheath  and  in 
contact  with  it  will  reduce  the  IR  drop  along  the 
sheath  and  potential  differences  within  the  cable. 
Other  remedial  measures  include: 
(a)   Increasing  core  to  sheath  dielectric  strength. 
This  method  is  of  major  importance  in  toll  cables, 
where  the  voltages  between  core  and  sheath  appear 
most  frequently,  but  for  exchange  cable,  where  the 
voltages  between  conductors  are  of  more  concern,  this 
method  is  of  lesser  importance. 

(6)   Employing  pole  protection  wires  on  exchange  •§  & 
cables  extending  into  rural  areas  where  cable  terminals 
are  infrequent. 

(c)  Bonding  to  power  system  common  neutrals  at 
frequent  intervals  in  built-up  areas  where  many  work- 
ing drops  are  connected  through  protector  blocks  to 
the  cable  sheath.  In  such  cases,  the  cable  sheath  be- 
comes closely  tied  in  with  the  neutral  through  the  drops,  station  protector  grounds,  and 
power  secondary  services.  By  proper  bonding  (at  about  ^-mile  intervals  or  less)  between 
sheath  and  neutral  (assuming  the  neutral  to  be  continuous  and  well  grounded),  the  neutral 
forms  a  parallel  path  with  the  cable  for  lightning  currents  and  provides  effective  shielding. 
Under  such  conditions,  cable  damage  from  power  contacts  or  lightning  tends  to  be  re- 
stricted to  a  section  between  the  two  bonds  immediately  adjacent  to  the  power  source. 
Bonding  to  the  neutral  also  lowers  the  sheath  impedance  and  thus  assists  in  prompt 
de-energization  of  the  power  circuit  in  the  event  of  power-circuit  contact  with  the  sheath. 
Frequent  bonding  to  the  neutral  is  useful  for  cable  noise  mitigation. 
Where  bonding  between  the  aerial  cable  and  power  neutral  is  objectionable,  with  respect 
to  corrosion  on  the  associated  underground  cable  plant  (as  where  drainage  is  used),  the 
bonded  aerial  sheath  may  be  isolated  from  the  underground  sheath  by  installing  an  insulat- 
ing joint  at  their  junction,  reliance  being  placed  in  the  bonding  for  the  aerial  sheath  pro- 
tective grounding. 

Experience  indicates  that  it  is  advisable,  in  lightning  affected  areas,  to  protect  all  work- 
ing conductors  at  any  terminal,  serving  any  open  wire  or  drop  loop  over  1/2  mile  in  length, 
this  length  being  based  principally  on  judgment. 

Some  companies  (not  Bell  System)  employ,  in  some  cases,  fuses  (usually  5-  or  7-amp 
rating)  in  conjunction  with  protector  blocks  to  form  a  unit-type  protector  which  is  usually 
assembled  in  multiples  and  mounted  in  a  terminal  housing  for  installation  at  the  junction 
•of  open-wire  or  drop  loops  and  the  cable.  The  protector  blocks  ground  the  open-wire  or 

drop  loop  when  lightning  or  excessive  power  circuit 
potentials  are  applied,  and  the  fuses  open  the  circuit 
when  the  current  coming  into  the  cable  over  the  loop 
exceeds  the  fuse  rating. 

TOLL  CABLE  PROTECTION  (against  lightning) 
requirements  are  similar  to  those  of  the  exchange 
plant  except  that  the  distances  are  greater,  partd<m- 
larly  between  grounds.  Experience  indicates  that 
the  rate  of  lightning  troubles  on  aerial  tott  cables 
apparently  does  not  differ  greatly  from  such  troubles 
in  buried  cables,  but  the  ease  of  locating  and  clearing 
these  troubles  in  aerial  cables  as  compared  to  under- 
ground cables  indicates  less  need  for  comparable 
remedial  measures  on  the  aerial  cables. 

Figure  10  shows  the  distribution  of  lightning 
stroke  crest  currents,  based  on  a  large  number  of 
measurements  in  the  ground  structures  of  power- 
transmission  lines,  the  crest  value  varying  over  a  wide  range.  Measurements  indicate 
that  the  crest  valite  is  reached  in  5  to  10  microseconds  and  that  it  decays  to  half  its  maxi- 
.mum  value  in  25  to  100  microseconds.  The  crest  value  and  decay  time  to  half  value  of  the 


KX) 

50 
4O 
3O 

\ 

^x. 

""^ 

^ 

^v 

*s 

^ 

\ 

10 
5 

Y 

^ 

2345          1O         203O50        TOO 
PER   CENT  OF  STROKES   WHTH   CURRENTS 
EXCEEDING  ORDiNATE 

FIG.     10.      Distribution     of     Lightning 

Stroke    Crest    Currents    (Courtesy    Bell 

System) 


10-62 


TRANSMISSION  CIRCUITS 


current,  rather  than  wave  front  steepness,  are  of  primary  importance  with  respect  to 
voltages  between  the  sheath  and  conductors  of  aerial  or  buried  toll  cable. 

At  a  point  remote  from  ground  connections,  the  surge  impedance  to  ground  of  an  aerial 
cable  sheath  (any  size)  is  about  200  ohms  (400  ohms  in  each  direction  from  the  lightning 
stroke) .  For  a  crest  current  of  20,000  amp  the  sheath-to-ground  voltage  would  be  4  million 
volts.  The  conductors  within  the  sheath  attain  about  the  same  potential  to  ground  as 
the  sheath,  owing  to  capacitive  and  inductive  coupling  between  the  sheath  and  core  con- 
ductors. However,  some  voltage  difference  does  exist  between  the  sheath  and  core,  due 
to  the  IR  drop  along  the  sheath,  resulting  from  the  flow  of  lightning  current  on  the  sheath. 
This  voltage  difference  is  greatest  at  the  stroke  point,  decreasing  as  the  distance  from  the 
stroke  point  increases.  The  higher  the  sheath  resistance,  the  greater  this  voltage  difference 
between  sheath  and  core  will  be  for  a  given  lightning  current. 

The  dielectric  strength  between  core  and  sheath,  for  normal-dielectric-strength  cables, 
is  for  surges  about  2000  volts.  For  high-dielectric-strength  cables  (having  an  extra  core 
wrap),  a  4000- volt  value  for  surges  may  be  assumed. 

After  an  initial  puncture  in  the  insulation  near  the  stroke  point,  other  punctures  will 
thus  usually  occur  some  distance  away  in  either  or  both  directions.  Such  puncturing  may 
or  may  not  cause  permanent  failures,  depending  on  the  current  through  the  fault. 

A  sheath-to-core  voltage  of  breakdown  magnitude  may  develop,  owing  to:  (1)  a  large 
sheath  current  for  a  relatively  short  distance,  (2)  a  smaller  sheath  current  over  a  longer 
distance,  or  (3)  a  combination  of  (1)  and  (2),  depending  on  the  ground  paths  from  the 
sheath  and  their  locations  with  respect  to  the  lightning  stroke.  Permanent  damage  is 
likely  to  result  from  the  large  current  transfer  through  the  punctures  in  condition  (1) 
but  is  not  so  likely  in  (2)  because  of  the  smaller  transfer  of  current  from  sheath  to  con- 
ductors under  this  condition. 

The  voltages  in  buried  cables  are  usually  due  to  large  sheath  currents  over  short  distances 
(less  than  x/2  mile),  so  that  permanent  damage  is  likely  at  puncture  points. 

The  protection  of  aerial  toll  cables  may  consist  of  (1)  pole  protection  wires  (wires  bonded 
to  the  sheath  and  extending  down  the  pole  to  the  pole  butt  or  to  a  point  near  the  ground 
line),  (2)  aerial  shield  wires,  (3)  buried  shield  wires,  (4)  high-dielectric-strength  cable  (with 
double  core  wrap),  (5)  shields  within  the  cable,  or  (6)  protector  blocks  at  open-wire  junc- 
tions or  out  from  the  junction  about  a  mile  on  the  open  wire. 

Pole  protection  wires  may  serve  not  only  to  protect  poles  against  splintering  but  also 
to  provide  protection  to  aerial  cables  in  lightning  exposures.  Data  show  that  for  such 
wires,  properly  spaced  along  an  aerial  cable  line,  the  voltage  between  sheath  and  core,  due 
to  a  stroke,  decreases  as  the  number  of  such  wires  increases  up  to  about  10  for  100  meter 
ohm  resistivity  and  up  to  about  20  for  1000  ohm  resistivity.  However,  in  uniform  ex- 
posures,-"the  effectiveness  of  these  wires  decreases  for  aerial  cables  having  both  toll  and 

exchange  conductors  as  the  interval  (of  the  order  of 
1  mile)  between  subscriber  drop  locations  decreases, 
owing  to  the  low-breakdown  path  from  sheath  to 
ground  over  the  drop  loops  which  substantially  short- 
circuits  the  pole  protection  wires. 

The  effectiveness  of  these  wires  may  be  lowered 
where  a  large  number  of  them  are  employed,  as  the 
result  of  current  distribution  along  the  sheath  to  such 
grounds  from  the  stroke  point  increasing  the  net 
sheath-to-eore  voltage.  By  using  proper  length  gaps 
in  the  wires,  the  number  of  conducting  grounds  may 
be  limited  to  an  optimum  value. 

Where  pole  protection  wires  are  not  adequate  to 
give  required  lightning  protection  to  aerial  cables, 


1.8  2.2 

CABLE    DIAMETER  Bsl  *NCH£$ 


^        **  aerial  or  buried  shield  wires  may  be  of  advantage. 

oi.-  u    -D    .L        r        *     •  i   The  reduction  in   core-to-sheath  voltage   by   aerial 
Shield    Factors    for    Aerial     ,  .  ,  ,      . ,  .  , i_j.j  ^  A1_  .i^tAx  _u 


FIG.    II. 
Shield  Wires  (Courtesy  Bell  System) 


shield  wires  which  are  bonded  to  the  sheath  at  each 
pole  is  normally  substantial,  as  shown  by  the  shield 
factors,  Fig.  11.  When  an  aerial  cable  is  provided  with  buried  shield  wires,  the  voltage 
from  a  direct  stroke  is  about  the  same  as  that  for  a  buried  cable  of  the  same  size  with 
similar  shield  wires. 

High-dielectric-strength  cables  should  generally  be  employed  in  new  installations.  The 
choice  of  this  type  of  cable  may  obviate  the  necessity  of  other  remedial  measures,  but  in 
any  event  its  higher  dielectric  strength  is  an  advantage  since,  if  the  strength  is  doubled, 
the  stroke  current  must  be  increased  3  times  to  puncture  the  insulation. 

Communication  plant  may  be  damaged  by  contact  with  high-voltage  wires  or  power 
distribution  circuits.  The  most  important  consideration  in  protecting  cables  from  power- 


CABLE  SHEATH  CORROSION  10-63 

wire  contact  is  to  so  locate,  construct,  and  maintain  the  communication,  and  power  circuit 
plant  that  contacts  will  not  occur. 

Aerial  cables  are  grounded  at  offices,  through  their  connection  to  the  underground  cable 
plant,  underground  dips,  and  private  cable  entrances,  as  well  as  other  frequent  grounding, 
such  as,  in  some  cases,  to  multigrounded  power  circuit  neutrals.  This  practice  contributes 
to  provide  a  low-impedance  ground  to  de-energize  the  power  circuit  promptly  in  case  of 
contact  with  the  cable  plant. 

27.  CABLE  SHEATH  CORROSION 

ELECTROLYSIS.  Electric  currents,  flowing  in  the  earth,  may  result  from  (1)  stray 
currents  from  d-c  street-car  (trolley)  or  electrified  railway  systems,  (2)  stray  currents  from 
commercial  d-c  power  distribution  systems,  (3)  differences  in  the  chemical  composition  of 
films  on  the  cable  sheath  at  different  locations,  (4)  differences  between  the  composition 
of  films  on  sheaths  of  different  cables  at  the  same  location,  (5)  differences  in  the  electrolyte 
at  different  places,  or  (6)  a  number  of  other  conditions.  These  currents  passing  through 
damp  ground  cause  chemical  changes  to  take  place  at  electrode  surfaces,  such  as  cable 
sheaths,  which  may  or  may  not  affect  the  electrodes. 

Cable  sheath  corrosion  may,  in  general,  be  considered  as  occurring  in: 

1.  Stray-current  areas. 

2.  Non-stray-current  areas. 

The  principal  causes  of  cable  sheath  corrosion  are: 

1.  Stray  current-anodic  action. 

2.  Stray  current-cathodic  action. 

3.  Localized  action. 

4.  Galvanic  action. 

5.  Chemical  attack. 

Stray-current  areas  are  commonly  designated  as  those  in  which  the  cable-to-earth 
potentials  and  sheath  currents  are  established  principally  by  currents  straying  from  the 
rails  of  d-c  transportation  systems  and  utilizing  other  paralleling  paths  of  relatively  low 
resistance  in  the  earth,  such  as  cable  sheath  or  public  water  piping. 

Where  the  stray  current  enters  the  sheath  the  cable  is  negative  (cathodic)  to  the  earthr 
and  where  it  leaves  the  sheath  the  cable  is  positive  (anodic)  to  the  earth. 

Non-stray-current  areas  are  those  in  which  corrosion  may  occur  from  other  than 
stray  currents.  Currents  in  these  areas  usually  result  from  potential  gradients  due  to 
such  causes  as  differential  aeration,  differential  electrolyte  concentrations,  non-uniformi- 
ties in  the  sheath  metal,  or  potential  differences  between  different  metals. 

Current  from  stray-current  areas  may  flow  in  non-stray  areas,  such  as  along  an  under- 
ground toll  cable  sheath  extending  between  cities  where  street  cars  operate.  Current, 
usually  considered  as  "non-stray"  current,  such  as  that  resulting  from  galvanic  potentials, 
may  also  be  present  in  stray-current  areas,  although  its  effect  is  largely  overshadowed  by 
the  stray  currents  in  these  areas. 

Anodic  corrosion,  due  to  stray  currents,  results  because: 

1.  The  metal  becomes  positively  ionized  and  goes  into  solution. 

Pb  -  2e  ->  Pb++     (e  =  electron) 

2.  The  anions  (ions  bearing  negative  charges  and  migrating  to  an  anode)  in  the  elec- 
trolyte are  attracted  to  and  contact  the  sheath,  where  they  lose  their  charge,  become 
chemically  active,  and  attack  the  lead  sheath. 

If  the  ions  are  chlorine 

Cl-  -  e  ->  Cl 
and 

Pb  +  2C1  -»  PbCl2     (lead  chloride) 

Note:  A  positive  ion,  in  the  case  of  Pb4""*",  is  an  atom  from  which  two  electrons  have  been  removed; 
a  negative  ion,  in  the  case  of  Cl"",  is  an  atom  to  which  one  extra  electron  has  been  added. 

In  anodic  action,  a  corrosion  product  does  not  always  adhere  to  the  sheath,  and  a  clean 
corroded  area  on  the  sheath  is  regarded  as  due  to  such  action.  The  corrosion  product,  if 
present  on  the  sheath,  is  usually  lead  chloride  or  lead  sulfate  or  both.  The  sheath  poten- 
tial, being  positive,  attracts  chloride  and  sulfate  ions  in  the  ground  water  to  the  sheath, 
and  these  ions  react  with  the  lead.  Under  severe  anodic  conditions  lead  peroxide,  PbOa, 
may  be  formed.  This  product  has  a  chocolate  color;  the  products  of  chloride,  sulfate,  and 
carbonate  of  lead  are  white. 


10-64  TRANSMISSION  CIRCUITS 

Cathodic  corrosion,  due  to  stray  currents,  though  not  as  yet  reproduced  in  the  laboratory, 
might  possibly  occur  when  the  cables  are  negative  to  earth  and  the  surrounding  earth 
contains  alkali  or  salts  of  sodium,  potassium,  calcium,  or  magnesium.  Hydrogen  ions  in 
the  electrolyte,  under  such  conditions,  are  attracted  to  the  sheath,  lose  their  charge,  and 
are  liberated,  resulting  in  a  decrease  in  hydrogen-ion  concentration  and  making  the  elec- 
trolyte alkaline.  The  alkali  forms  most  rapidly  and  with  greatest  concentration  at  points 
of  maximum  cathodic  current  density.  The  attack  on  the  sheath  is,  therefore,  not  uniform 
and  usually  affects  only  a  relatively  small  part  of  the  sheath. 

La  the  case  of  sodium  salts,  the  reaction  with  the  lead  is: 

H+  -f  e  -+  H 
Na+  +  OH~  — >  NaOH     (sodium  hydroxide) 

The  sodium  hydroxide  forming  at  the  sheath  dissolves  the  lead,  and  the  final  reaction 
usually  results  in  the  formation  principally  of  lead  monoxide,  PbOT  lead  carbonate,  and 
sodium  carbonate.  The  sheath  pitting  may  be  similar  to  that  caused  by  anodic  action, 
but  the  cathodic  corrosion  is  characterized  by  the  usually  bright  orange-red  color  of  the 
lead  monoxide. 

Galvanic  action  usually  results  when  two  dissimilar  metals  are  in  electric  contact  with 
an  electrolyte  and  also  are  metallically  connected.  Where  cables  and  bare  copper  or  gal- 
vanized cable  rack  supports  are  present  in  a  flooded  manhole,  galvanic  action  may  occur. 
Lead  is  anodic  to  copper  and  cathodic  to  the  zinc  galvanizing,  the  anode  corroding  in  each 
case.  Corrosion  may  appear,  in  the  presence  of  moisture,  on  wiped  joints  and  soldered 
seams  on  cable  sleeves,  where  the  solder  may  be  anodic  to  the  lead. 

When  two  dissimilar  metals  in  an  electrolyte  are  connected  metallically,  the  metal  with 
the  higher  solution  pressure  becomes  the  anode  and  corrodes.  Current  passes  from  the 
anode  through  the  electrolyte  to  the  cathode  and  completes  the  circuit  to  the  anode  through 
the  metallic  connection. 

In  alkaline  solutions,  lead  tends  to  become  electronegative  and  may  be  anodic  with 
respect  to  another  metal  such  as  iron.  Thus,  lead  corrosion  may  result  where  cable  is 
installed  in  iron  conduit  and  such  solutions  are  present.  Under  these  conditions  current 
may  pass  from  the  lead  through  the  solution  to  the  iron  and  return  to  the  lead  at  points 
of  metallic  contact  between  the  iron  conduit  and  the  sheath. 

Local  action  between  adjacent  areas  of  the  sheath  may  result  from  variations  in  sheath 
material,  such  as  foreign  particles  in  the  lead,  and  from  differences  in  surface  conditions 
due  to  abrasion.  Corrosion  from  the  action,  under  anodic  conditions,  may  occur  in  spots 
rather  than  uniformly  over  the  sheath  surface,  and  if  the  action  continues  for  a  period  of 
years  sharply  defined  pits  may  appear  in  the  sheath. 

Concentration  cells  may  be  formed  by  a  change  in  the  concentration  of  the  salts  in  an 
electrolyte,  causing  corresponding  changes  in  the  potential  of  a  given  electrode  in  contact 
with  the  electrolyte;  or  by  two  electrolytes,  with  equal  concentration  of  different  salts, 
producing  different  electrode  potentials  on  the  same  metal.  Such  a  cell  might  be  established 
by  waste  products  from  a  factory  or  sewer  entering  a  sloping  conduit  (with  cable)  at  a 
conduit  joint  and  concentrating  along  a  section  of  sheath.  The  cell  might  then  result  be- 
tween the  section  of  sheath  having  the  concentration  and  an  adjacent  section  not  affected 
by  the  waste. 

Differential  aeration,  which  may  be  considered  a  special  form  of  concentration  cell, 
results  in  sheath  corrosion,  owing  to  a  variation  in  the  concentration  of  dissolved  oxygen 
in  the  electrolyte  in  contact  with  the  sheath.  Where  a  sheath  is  continually  wet  by  water 
dripping  or  condensing  on  it,  there  will  be  more  oxygen  in  the  moisture  exposed  to  the  air 
than  where  the  moisture  is  shielded  by  an  absorbent  material  such  as  a  layer  of  silt  on  the 
sheath.  The  cable  sheath  in  contact  with  the  electrolyte  with  a  deficiency  of  oxygen  will 
be  anodic  and  subject  to  corrosion. 

Battery  action  in  the  soil  might  result  in  sheath  corrosion  where  the  sheath  was  not  one 
of  the  cell  electrodes  but  acted  as  a  conductor  of  the  local  current.  As  an  example,  if  an 
iron  pipe  passed  through  a  bed  of  cinders  which  acted  as  a  carbon  electrode,  and  the 
adjacent  section  of  pipe  was  in  contact  with  ordinary  soil,  a  cell  would  be  established  with 
the  carbon  electrode  negative  to  the  adjacent  section  of  pipe.  Part  of  the  resulting  current 
from  this  section  might  enter  a  nearby  paralleling  cable  sheath  and  leave  the  sheath  near 
tEe  cinders,  at  which  point  sheath  corrosion  might  result. 

Protective  films  formed  over  the  sheath  by  natural  processes  are  usually  helpful  in  pre- 
venting sheath  corrosion.  Films  formed  from  silicates  are  generally  continuous,  adherent, 
insoluble,  and  helpful,  whereas  films  resulting  from  nitrates  in  the  soil  tend  to  prevent  film 
aid  corrosion. 


CABLE  SHEATH  CORROSION  10-65 

REMEDIAL  MEASURES  for  controlling  corrosion  are  varied  and  often  complex,  be- 
cause the  problems  of  corrosion  may  differ  over  a  wide  range  of  causes  and  conditions  and 
extend  over  relatively  large  areas. 

Generally,  the  first  considerations  in  the  control  of  cable  sheath  corrosion  are  electrical, 
involving  a  thorough  study  of  electrical  conditions  affecting  the  cable  plant.  Chemical 
control  can  sometimes  be  used  successfully  where  electrical  methods  are  not  practicable. 
The  general  attack  on  sheath  corrosion  problems  by  electrical  methods  consists  of: 

1.  Limiting  current  entering  the  sheath  to  the  extent  practicable. 

2.  Providing  metallic  paths  by  which  current  entering  the  sheath  may  leave  it  without 
damage  to  the  sheath  or  other  metallic  structures. 

Protective  arrangements  of  any  nature  should  be  such  as  to  minimize  the  probability 
of  impressing  current  on  other  underground  plants,  either  privately  or  publicly  owned. 

The  limitation  of  current  pick-up  by  a  given  underground  cable  sheath  requires  that  the 
sheath  be  kept  free  of  all  connections  to  other  grounded  metallic  structures  except  those 
connections  specified  as  part  of  the  general  remedial  plan,  and  that  the  sheath  should  not 
be  made  more  negative  to  earth  in  any  area  than  is  necessary  under  practical  design. 

The  cable  sheath  should  be  maintained  at  slight  negative  potential  to  earth  to  (1)  limit 
current  pick-up  which  increases  with  increase  in  negative  potential,  (2)  lessen  the  possi- 
bility of  large  positive  pipe-to-cable  voltages,  and  (3)  reduce  the  chances  for  cathodic 
corrosion. 

The  increasing  use  of  bonding  between  aerial  cables  and  multigrounded  power  neutrals 
for  noise  induction  and  protection  reasons  increases  the  tendency  to  discharge  current  to 
the  underground  cable  plant.  Insulating  joints  may  be  used  at  underground-aerial  cable 
junctions,  where  required,  and  electrolytic  capacitors  may  be  employed  to  bridge  the 
insulating  joint  if  capacitors  are  necessary  for  noise  induction  or  protection  purposes. 

Cathodic  corrosion  of  sheaths  may  occur  where  the  cable-to-earth  potential  exceeds  a 
few  tenths  of  a  volt  negative.  In  the  presence  of  salts  this  potential  may  not  be  over  0.2 
volt;  under  other  conditions  cathodic  corrosion  will  not  result  at  any  potentials  encoun- 
tered, where  an  effective  drainage  system  exists. 

For  this  type  of  corrosion,  it  is  essential  to  maintain  low  negative  cable-to-earth  poten- 
tials by  using  (1)  an  adequate  drainage  system,  (2)  insulating  joints  shunted  by  resistors 
if  necessary,  (3)  corrosion-protected  cables  for  replacement  of  cables  which  fail,  (4)  periodic 
flushing  of  ducts  which  accumulate  alkali,  or  (5)  reverse  drainage  employing  controlled 
currents  to  the  cables  by  connecting  to  positive  points  on  rail  systems  or  to  pipes  (expe- 
rience limited). 

Other  underground  structures,  such  as  public  water  or  gas  piping,  are  also  drained  in 
many  areas.  A  metallic  piping  system  in  direct  contact  with  the  earth  usually  has  a  very 
low  leakage  resistance,  and  any  reduction  in  its  potential  by  drainage  tends  to  lower  the 
earth  potential  in  its  immediate  vicinity.  Where  the  earth  potential  is  thus  lowered,  the 
current  discharge  from  nearby  telephone  cables  tends  to  increase.  This  condition  may 
complicate  the  telephone  cable  drainage  scheme,  but  generally,  owing  to  the  low  leakage 
resistance  of  the  piping  system,  the  adverse  effects  are  not  extensive.  A  coordinated 
drainage  plan  for  all  underground  systems  is  usually  necessary  where  two  or  more  systems 
serve  the  same  general  area  affected  by  stray  currents. 

Anodic  corrosion,  which  is  the  only  type  of  corrosion  that  has  been  found  so  far  in  non- 
stray  areas,  results  from  current  leaving  the  cable  sheath  for  any  reason.  This  type  of 
corrosion  may  be  due  to:  (1)  currents  flowing  from  the  cable  into  the  electrolyte  in  the 
duct  and  back  to  the  cable  without  leaving  the  duct,  as  a  result  of  differential  aeration, 
differential  electrolyte  concentration,  or  non-uniformity  of  the  sheath  composition  (though 
this  type  of  corrosion  is  local,  many  such  local  cells  may  exist  along  a  cable  and  cause 
corrosion  over  a  long  section  of  the  cable) ;  and  (2)  currents  flowing  from  the  cable  to  the 
electrolyte  in  the  duet,  thence  to  the  earth  outside  of  the  duct,  and  finally  returning  to  the 
cable  at  some  relatively  remote  point.  In  (2)  the  driving  electromotive  forces  may  result 
from  the  same  causes  as  given  in  (1)  above,  but  the  cells  are  materially  lengthened  and 
the  currents  are  known  as  long  ceU  currents.  Such  electromotive  forces  may  result  from 
potential  gradients  in  the  earth  due  to  currents  associated  with  corrosion  cells  on  long 
paralleling  piping  or  to  other  natural  causes,  including  magnetic  storms;  or  they  may 
result  from  a  potential  established  as  a  result  of  sheath  contact  with  other  metal,  such  as 
copper  or  iron  piping;  or  they  may  result  from  remote  railway  currents  flowing  into  the 
non-stray  area. 

TESTING  METHODS  AND  MITIGATIVE  MEASURES,  as  employed  in  non-stray- 
current  areas,  areT  in  general,  somewhat  similar  to  those  employed  in  stray-ctirrent  areas, 
but  various  testing  refinements  are  usually  required  in  the  non-stray  areas.  For  example, 
in  determining  the  IR  drop  or  direction  of  current  flow  in  the  earth  as  it  may  affect  under- 


10-66  TRANSMISSION  CIRCUITS 

ground  cable  sheath,  it  is  necessary  to  employ  electrodes,  for  contacting  the  earth,  which 
are  identical  in  potential  or  differ  by  a  known  and  constant  potential.  Such  an  electrode, 
designated  a  half-cell,  consists  of  a  non-conducting  container  enclosing  a  metallic  electrode 
suspended  in  an  electrolyte.  The  electrolyte  fills  a  porous  cap,  which  forms  the  bottom 
of  the  container  and  through  which  the  cell  makes  contact  with  the  earth.  This  cell  makes 
use  of  the  fact  that  the  potential  difference  between  electrolytes  varies  over  a  very  small 
range  as  compared  to  the  potential  difference  between  metals  when  used  as  electrodes. 

Experience  has  shown  that  corrosion  is  not  usually  a  problem  with  respect  to  buried 
jute-protected,  tape-armored,  and  thermoplastic-covered  cables  in  non-stray  areas. 
Electrical  tests  on  such  cables  in  non-stray  areas  are  not,  in  general,  considered  necessary. 

Forced  drainage  has  been  successfully  employed  in  anodic  areas  in  the  protection  of 
underground  cable  plant  in  non-stray  areas.  Random  contacts  between  telephone  cables 
and  other  metallic  structures,  such  as  water  piping  and  steel  buildings,  usually  interfere 
with  effective  drainage  of  the  cables  and  should  be  eliminated,  although  other  considera- 
tions, particularly  noise  induction  and  protection,  may  impose  numerous  difficulties  and 
problems  in  such  eliminations. 

Forced  drainage  requires  a  separate  d-c  source  of  potential,  such  as  a  rectifier  (commonly 
used)  or  a  battery,  with  the  negative  terminal  connected  to  the  cable  sheath  and  the  posi- 
tive terminal  connected  to  a  negative  bus  or  special  ground.  Current  is  forced  from  the 
sheath  to  the  bus  or  ground. 

Rectifiers  available  for  drainage  purposes  have  suitable  d-c  voltage  and  current  outputs 
and  usually  operate  from  either  115-  or  230-volt  a-c  commercial  supply.  They  are  made 
in  several  types,  including  dry-disk  and  tube  types,  by  a  number  of  electrical  equipment 
manufacturers. 

Galvanic  anodes  requiring  no  external  pwer  supply  may  provide  the  required  amount 
of  forced  drainage  under  favorable  conditions  when  buried  in  the  ground  and  connected 
to  the  sheath  by  copper  wire.  The  anodes  may  consist  of  a  metal  negative  to  lead,  such  as 
zinc,  aluminum,  or  magnesium.  Magnesium  appears  favorable,  because  of  its  relatively 
high  negative  potential  (about  1  volt)  to  lead.  The  anode,  being  buried  in  the  soil  and 
discharging  current,  will  gradually  be  consumed. 

To  be  effective,  the  anode  must  have  a  low  resistance  to  ground,  and  this  depends  on  its 
shape  and  the  surrounding  earth  resistivity.  For  a  cylinder  4  in.  in  diameter  by  20  in. 
long,  the  resistance  to  ground  (without  special  environment)  is  about  equal  to  the  earth 
resistivity  in  meter  ohms. 

Chemical  attack  usually  requires  an  analysis  of  the  corrosion  products  and  a  determina- 
tion of  the  source  of  the  chemical  attacking  the  cable  sheath  in  order  to  apply  suitable 
remedial  measures.  Chemists  may  be  of  assistance,  in  difficult  cases,  in  determining  the 
nature  of  the  attacking  chemicals.  Lead  monoxide  corrosion  is  indicative  of  cathodic  action 
or  alkali  attack.  Very  few  cases  show  a  definite  single  cause  of  corrosion.  The  previous 
history  of  corrosion  of  the  affected  cable  may  be  of  value  in  arriving  at  the  causes  of  the 
corrosion. 

Alternating  current,  principally  because  of  its  rapid  and  equal  reversals  of  potential 
between  positive  and  negative  values,  is  not  considered  an  important  cause  of  cable  sheath 
corrosion. 

Corrosion-protected  cable  for  installation  in  underground  conduit  can  be  made  available 
with  the  same  core  make-up  as  the  plain  lead-covered  cable  with  which  it  may  be  asso- 
ciated. This  protected  cable  may  be  useful  where  the  lead  sheath  would  be  subject  to 
corrosive  action  without  the  protection  and  where  it  is  more  attractive  than  other  remedial 
measures.  This  type  of  cable  may  be  employed  in  situations  such  as  (1)  near  chemical 
plants  or  other  locations  where  chemical  attack  has  been  experienced  or  may  occur,  (2) 
where  alkaline  attack  (cathodic  action)  might  develop,  or  (3)  where  corrosion  has  occurred 
in  subsurface  dips. 

One  type  of  protection  consists  of  two  reversed  layers  of  Sisalkraft  paper  and  an  outer 
layer  of  rubber-filled  tape,  the  sheath  and  each  layer  being  flooded  with  an  asphalt  com- 
pound. A  non-adhesive  coating  is  applied  on  the  outside  covering  to  prevent  sticking  in 
handling.  The  protection  increases  the  cable  diameter  about  0.2  in, 

Suitable  alarms  and  pilot  wires  to  a  centralized  maintenance  center  may  be  employed, 
when  facilities  are  available  and  as  required,  to  indicate  critical  changes  in  potentials  and 
current  flow  affecting  the  cable  plant. 


FOREIGN  WIRE  RELATIONS  10-67 

COORDINATION  OF  COMMUNICATION 
AND  POWER  SYSTEMS 

By  John  D.  Taylor  and  Howard  L.  Davis,  Jr. 

28.  FOREIGN  WIRE  RELATIONS 

In  order  to  insure  safety  to  persons  and  property,  economy  of  operation,  and  good 
service,  in  areas  served  by  both  overhead  communication  and  power  systems,  it  became 
evident,  as  the  systems  began  to  expand,  that  the  companies  involved  should  establish 
and  follow  a  plan  of  cooperation  in  the  construction  and  operation  of  their  respective 
plants.  For  a  number  of  years  individual  cooperative  efforts  were  carried  on,  but  the  spe- 
cific solutions  of  the  problems  that  developed  were  not  applicable  in  a  general  way. 

Early  in  1921  steps  were  taken  by  both  interests  on  a  nationwide  scale  to  formulate  a 
basis  of  common  understanding  and  to  establish  permanent  joint  committees  and  sub- 
committees for  study  and  recommendations  relating  to  mutual  problems.  As  a  result  of 
continuous  study  and  research  by  these  subcommittees  and  sponsors,  the  Joint  General 
Committee  of  the  Edison  Electric  Institute  and  Bell  Telephone  System  prepared,  and  the 
representative  interests  approved,  several  general  reports,  of  which  the  following  are  the 
principal  ones  in  effect  today: 

1.  Principles  and  Practices  for  the  Inductive  Coordination  of  Supply  and  Communication 
Systems,  Dec.  9,  1922.* 

2.  Principles  and  Practices  for  the  Joint  Use  of  Wood  Poles  by  Supply  and  Communication 
Companies,  Feb.  15,  1926.* 

3.  Inductive    Coordination — Allocation  of   Costs   between   Supply  and    Communication 
Companies,  Oct.  15,  1926.* 

In  general,  the  Principles  and  Practices  provide,  in  addition  to  other  important  items, 
that 

(a)  All  supply  and  communication  circuits  with  their  associated  apparatus  should  be 
located,  constructed,  operated,  and  maintained  in  conformity  with  general  coordinated 
methods  based  on  the  concept  of  rendering  either  service  without  interference. 

(6)  Where  general  coordinated  methods  will  be  insufficient,  suitable  specific  coordinated 
methods  should  be  applied,  most  conveniently  and  economically,  to  prevent  interference 
with  either  service,  present  and  known  future  factors  being  taken  into  account. 

(c)  The  companies  serving  any  given  area  should  fully  cooperate  with  each  other  in 
carrying  out  the  accepted  principles,  based  on  arriving  at  the  best  engineering  solution  of 
each  situation,  as  it  arises,  for  all  the  companies  involved. 

(d)  Where  conditions  and  the  nature  of  the  supply  and  communication  circuits  permit, 
joint  use  of  poles  (particularly  in  urban  areas)  is  generally  preferable  to  separate  lines, 
when  justified  by  considerations  of  safety,  economy,  and  convenience,  and  assuming  that  a 
satisfactory  agreement  is  reached  between  the  parties  concerned. 

(e}  When  supply  and  communication  facilities  occupy  the  same  section  of  highway  and 
joint  use  is  not  desirable,  each  type  of  facility  should  be  confined  to  one  side  of  the  high- 
way, as  far  as  practicable,  thus  avoiding  unnecessary  crossings  and  expensive  guying. 

(/)  In  the  design,  construction,  and  operation  of  supply  and  communication  circuits 
and  equipment,  all  factors  contributing  to  inductive  influence,  inductive  couplings,  or 
inductive  susceptiveness  under  normal  or  abnormal  operating  conditions  should  be  limited 
to  the  extent  necessary  and  practicable. 

(g)  Each  utility  shall  be  the  judge  of  the  quality  and  requirements  of  its  own  service 
and  the  type  and  design  of  its  own  facilities. 

(h)  Coordination  costs  in  any  given  situation  of  proximity,  assuming  that  satisfactory 
results  have  been  attained  under  the  best  engineering  solution,  will  generally  be  allocated, 
so  that  each  company  involved  bears  its  equitable  portion,  including  its  own  betterments. 

The  basis  for  cooperation,  as  set  forth  in  detail  in  the  Principles  and  Practices,  has  con- 
tributed immeasurably  to  the  excellent  foreign  wire  relations  existing  among  the  many 
•wire-using  companies  who  supply  electric  and  communication  services  throughout  the 
country. 

The  cooperative  plan  was  later  extended  to  provide  for  a  Joint  General  Committee  of 
the  American  Railway  Association  (now  Association  of  American  Railroads)  and  Bell 
Telephone  System  in  1929,  a  Joint  General  Committee  of  the  Edison  Electric  Institute 

*  These  reports  are  now  combined  as  Reports  of  Joint  General  Committee  of  Edison  Electric  Institute 
and  Bell  Telephone  System  on  Physical  Relations  between  Electrical  Supply  and  Communication  Systems, 
reissued  July  1945. 


10-68  TRANSMISSION  CIRCUITS 

and  Western  Union  Telegraph  Company  in  1935,  and  a  Joint  General  Committee  of  the 
Association  of  American  Railroads  and  Edison  Electric  Institute. 

When  power  and  communication  facilities  serve  the  same  areas  and  are  supported  on 
overhead  structures  and  where  the  two  types  of  facilities  are  in  close  physical  relation  or 
inductively  coupled  or  both,  situations  of  proximity  generally  are  unavoidable.  Communi- 
cation facilities,  operating  at  relatively  low  voltages  and  currents,  are  not  designed  to 
withstand  the  normal  or  abnormal  power  circuit  voltages  and  currents  which  may  be 
impressed  upon  them  by  direct  contact  or,  in  severe  influences  or  couplings,  by  induction. 
In  some  cases  of  contact  or  induction,  the  service  may  be  only  interrupted  or  degraded, 
but  in  more  severe  situations  the  service  may  be  interrupted  and  the  facilities  damaged 
with  or  without  personnel  hazard. 

It  is  thus  imperative  that  the  Principles  and  Practices  be  closely  adhered  to,  in  order 
that  both  services  may  be  furnished  the  public  in  a  safe,  economical,  and  satisfactory 
manner. 

The  design  and  application  of  coordinative  measures  by  the  wire-using  companies  in- 
volve two  broad  fields  of  coordination,  structural  and  inductive.  Both  fields  have  been 
for  a  number  of  years  and  will  continue  to  be  under  intensive  study,  directed  toward  im- 
proving methods  and  securing  further  economies  without  sacrificing  but,  when  practicable, 
bettering  present  safety  and  service. 

29.  STRUCTURAL  COORDINATION 

Structural  coordination,  as  the  name  implies,  consists  of  planning,  designing,  construct- 
ing, and  maintaining  the  physical  overhead  plant  of  each  company,  with  due  consideration 
for  the  plans  and  plant  of  all  other  companies  involved,  so  that  safety  and  overall  economy 
will  be  attained. 

SITUATIONS  OF  PROXIMITY  are  created  when  power  and  communication  lines  are 
so  located  with  respect  to  each  other  as  to  parallel,  cross-over,  occupy  the  same  poles 
(joint  use),  or  require  consideration  of  line  wire,  guy,  or  pole-mounted  equipment  clear- 
ances. The  overbuilding  of  one  type  of  service  by  the  other  is  considered  objectionable, 
joint  use  usually  being  the  better  solution,  with  the  power  wires  in  the  upper  position  in 
all  cases  of  proximity. 

The  National  Electrical  Safety  Code,  Fifth  Edition,  Part  2  ( National  Bureau  of  Standards 
Handbook  H32),  issued  Sept.  23,  1941  (hereinafter  referred  to  as  the  Code),  which  has 
been  approved  by  the  American  Standards  Association,  presents  Safety  Rules  for  the 
Installation  and  Maintenance  of  Electric  Supply  and  Communication  Lines.  These  rules 
embody  specific  minimum  requirements,  and,  though  not  complete,  they  are  intended  to 
cover  those  points  which  are  most  important  for  the  safety  of  employees  and  the  public. 
This  code  is  acceptable  to  the  various  wire-using  organizations  throughout  the  country 
and,  except  as  modified  by  more  exacting  state  or  local  regulations,  is  applied,  together 
with  the  Principles  and  Practices  discussed  previously,  generally  throughout  the  country. 
Where  a  certain  coordinative  problem  arises,  not  specifically  covered  in  the  Code,  the 
companies  involved  agree  on  the  best  engineering  solution  for  the  problem.  The  Code 
also  provides  for  modifying  or  waiving  its  requirements  in  any  given  case  where  such 
requirements  are  inapplicable,  not  justified,  or  impracticable,  or  where  equivalent  or  safer 
construction  can  be  more  readily  provided  by  other  means.  It  is,  therefore,  a  practicable 
and  flexible  guide. 

Structural  requirements  and  clearances  between  electric,  railway,  and  communication 
structures,  wires,  and  equipment  must  be  adequately  provided  for,  as  set  forth  in  the  Code. 
Owing  to  the  large  amount  of  detailed  information  necessary  in  specifying  these  require- 
ments under  the  numerous  conditions  encountered  in  practice,  they  will  not  be  given  here, 
but  they  may  readily  be  obtained  from  the  Code. 

Certain  fundamental  concepts  (other  than  those  previously  enumerated  under  Principles 
and  Practices)  in  this  work  are  generally  accepted  as  good  engineering  practice  and  may 
be  stated  in  general  terms: 

(a)  The  mechanical  design  and  construction  of  electric  (supply)  and  comrni.mTCfl.tion 
systems  should  conform  to  good  modern  practice. 

(b)  When  changes  are  made  in  systems  or  methods  of  operation,  consideration  should 
be  given  to  decreasing  inductive  influences  and  susceptiveness,  when  practicable. 

(c)  Coordinated  systems  should  be  maintained,  so  that  abnormal  conditions  affecting 
either  service  will  be  minimized  and  prompt  action  will  be  taken  to  eliminate  such  condi- 
tions when  they  do  occur. 


STRUCTURAL  COORDINATION 


10-69 


(tf)  Supply,  communication,  and  trolley  circuits  should  occupy  levels  in  the  order 
named,  with  supply  circuits  at  the  top  level.  Also,  where  supply  lines  carry  different 
voltages,  the  higher-voltage  lines  are  usually  placed  above  those  of  lower  voltage. 

(e)  Joint  use  should  be  considered  when  it  can  be  employed  with  reasonable  safety  and 
convenience,  economically,  and  without  appreciable  service  detriment. 

JOINT  USE  of  poles  is  a  very  desirable  means  of  coordinating  supply  and  communi- 
cation facilities  where  this  type  of  construction  is  feasible.  Various  types  of  construction 
are  employed,  depending  upon  the  types  of  facilities  involved,  but  in  any  case  the  con- 
struction is  designed  to  conform  to  the  latest  practices  and  safeguards. 


,3    3    3 


ft 


3 


3    9 


SUPPLY  ATTACHMENTS,  EXCEPT  VER-  N 
TICAL  RUNS,  STREET  LAMPS,  SPAN 
WIRES,  TROLLEY  CONTACT  CONDUC- 
TORS AND  ASSOCIATED  FEEDERS,  SHALL 
PREFERABLY  BE  ABOVE  COMMUNICA- 
TION ATTACHMENTS  (PART  1- PAR, 2-00 


8     8     8J 


a   a    a 


_i 


COMMUNICATION  ATTACHMENTS.  EX- 
CEPT VERTICAL  RUNS,  SHALL  PREFER- 
ABLY BE  ABOVE  TROLLEY  CONTACT 
CONDUCTORS  AND  ASSOCIATED  FEED- 
ERS   (  PARTI -FAR.  Z.Ot) 


TROLLEY  CROSSARMS  CARRYING  DC 
FEEDERS  Of  NOT  OVER  750  VOLTS 


NOTE. 

REFERENCES,  SUCH  AS   (PART  1-PAFL  2.01)  RE- 
FER TO  THE  TEXT  IN  A  REPORT  OF  THE  JOINT 
COMMITTEE  ON  PLANT  COORDINATION  OF  THE 
EDISON  ELECTRIC  INSTITUTE  AND  THE  BELL 
TELEPHONE  SYSTEM,  ENT1TLEO  *  JOINT  POLE 
PRACTICES  FOR  SUPPLY  AND  COMMUNICATION 
ORCmTS,"  ISSUED  OCTOBER  2&,1945 


VOLTAGE  OF  SUPPLY 
CIRCUIT  CONCERNED 

DIMEN- 
SION 

MINIMUM    CROSSARM 
SPACIWt  IN  INCHES 

DIMEN- 
SION 

MINIMUM  SEPARATION  BETWEEN 
CONDUCTORS  IN  INCHES 

0-8,700 

A 

48 

a 

•40 

OVER  -S,700 

72 

«0 

(PART  2-  PAR.  1O.O3} 

FIG.  1.    Relative  Position  of  Attachments,  Showing  Vertical  Clearances  and  Climbing  Space  (Joint 
Pole  Practices  for  Supply  and  Communication  Circuits,  Oct.  29,  1945} 

For  joint-use  construction,  commonly  employed  in  urban  areas  for  local  distribution, 
Figs.  1,  2,  3,  and  4  show  the  usual  typical  construction  features  aini  clearances  eonsidered 
good  modern  practice  and  meeting  Code  requirements.  Adequate  clearances  are  essential 
for  the  protection  of  personnel  and  property. 

Normal  joint-use  construction  is  applicable  to  construction  involving  comrnumoaztion 
cables  or  conductors  and  supply  cables  or  conductors  of  the  following  types: 

(a)  Constant-potential  a-c  supply  circuits  normally  operating  at  voltages  between  750 
and  5000  volts  between  conductors  and  not  over  2900  volts  to  aeutral  or  ground. 

(6)  Constant-current  supply  circuits  of  not  more  than  7.5  amp  regardless  of  the  voltage, 
and  of  more  than  7.5  amp  where  the  open-circuit  voltage  of  the  supply  transformer  is  not 
more  than  2§00  volts. 

(c)  Constant-potential  a-c  supply  circuits  normally  operating  at  more  than  5000  volts 
between  conductors  or  more  than  2900  volts  to  neutral  or  ground,,  and  eonstaiit-ourreat 


10-70 


TRANSMISSION  CIRCUITS 


SUPPLY  TRANSFORMERS  SHALL  BE  LOCATED  / 

ABOVE   COMMUNICATION  ATTACHMENTS  EX-^"1  i 

CEPT   WHERE  IMPRACTICABLE  BECAUSE  Of  J 

THEIR   SIZE  OR  WEIGHT  (PART  1-  PAR.  2.02)  } 


COMMUNICATION    CABLE- 


SUPPLY  GROUNDING   CONDUCTORS  SHALL  BE  PROVIDED 
WITH  AN   INSULATING  COVERING   FROM   THEIR  LOWEST 
POINT  UP  TO  AT    LEAST  4O  INCHES  ABOVE  THE   HIGHEST 
COMMUNICATION  OR  TROLLEY  ATTACHMENT,  EXCEPT 
THAT  THIS  COVERJNG  NEED   NOT  EXTEND  BELOW  THE 
TOP  OF  THE  PROTECTION  PROVIDED  FOR   8  FEET  ABOVE  — 
GROUND  AND  MAY  BE  OMITTED,  IF  THERE  ARE  NO  TROL- 
LEY ATTACHMENTS,  FROM  GROUNDING  CONDUCTORS 
WHJCH  ARE  METALLICALLY  CONNECTED  TO  A  CONDUCTOR 
WHICH  FORMS  A  PART  OF  AN  EFFECTIVE  GROUNDING 
SYSTEM      (PART  2- PAR.  11.04  AND  11,07) 


SUPPLY   GROUNDING  CONDUCTOR 


DIMENSION 

TRANSFORMER 
PRIMARY  VOLTAGE 

MINIMUM    SEPARATION 
IN   INCHES 

A 

0-8,700 

4-0 

OVER   Q,70O 

60* 

(PART  2-PAR.  10.03) 
#  TRANSFORMER   CASES,  Oft   OTHER  APPURTENANCES,  IF 
EFFECTIVELY  GROUNDED*  MAY  HAVE  A  SEPARATION  NOT 
LESS   THAN   4O   INCHES  FROM  COMMUNICATION  ATTACH- 
MENTS 

NOTE: 

REFERENCES, SUCH  AS  (PART  2-PAR.  10.03)  REFER  TO  THE 
TEXT  IN  A  REPORT  OF  THE  JOINT  COMMITTEE  ON  PLANT 
COORDINATION  OF  THE  EDISON  ELECTRIC  INSTITUTE  AND 
THE  BELL  TELEPHONE  SYSTEM,  ENTITLED  "JOINT  POLE 
PRACTICES  FOR  SUPPLY  AND    COMMUNICATION  CIRCUITS," 
•SSUED  OCTOBER  29,1945 


,THIS   PROTECTIVE  COVERING 
MAY  BE  OMITTED  FOR  GROUND- 
ING  CONDUCTORS  WHICH: 
1.  ARE  METALLICALLY  CONNECT- 
ED TO  A  CONDUCTOR  WHJCH 
FORMS  A  PART  OF  AN  EFFEC- 
TIVE GROUNDING  SYSTEM, OR 
2.IN  RURAL  DISTRICTS,  HAVE  A 
WEATHER-RESISTANT  COVgR- 
ING    (PART  2-PAR. 11.03) 


AT  LEAST  6  FEET 
(PART  2-  AR.tl.Q3) 


FIG.  2.     Supply  Transformer  Installation,  Showing  the  Separation  from  Communication  Cables  and 
Conductors  (Joint  Pole  Practices  for  Supply  and  Communication  Circuits,  Oct.  29,  1945) 


supply  circuits  of  more  than  7.5  amp  where  the  open-circuit  voltage  of  the  supply  trans- 
former is  more  than  2900  volts,  provided  that : 

1.  The  supply  and  communication  circuits  are  so  constructed,  operated,  and  maintained 
that  the  supply  circuits  will  be  promptly  de-energized,  both  initially  and  following  subse- 
quent breaker  operations,  in  the  event  of  contact  with  the  communication  plant. 

2.  The  voltage  and  current  impressed  on  the  communication  plant,  in  the  event  of  a 
contact  with  the  supply  conductors,  are  not  in  excess  of  the  safe  operating  limit  of  the 
communication  protective  devices. 


(d)  Any  effectively  grounded  supply  cables,  located  above  communication  cables  or 
conductors  or  carried  on  effectively  grounded  suspension  strand,  where  the  supply  voltage, 
between  conductors  is  more  than  750  volts. 


STRUCTUEAL  COORDINATION 


10-71 


VERTICAL  SUPPLY  CABLE 
OR  CONDUCTORS  COVERED 

OR  ENCLOSED 
(PART  2-PAR.  11.04) 


COMMUNICATION  GROUNDING 

CONDUCTOR  COVERED  WITH 

WOOD  MOLDING 
(PART  2- PAR.  11.10) 


SUPPLY  GROUNDING 
CONDUCTOR  COVERED 
WITH  WOOD  MOLDING 
(PART  2-PAR.  JU>7) 


APPOXtMATELY  5  INCHES 


•VERTICAL  COMMUNICATION 
CABLE  COVERED  OR  ENCLOSES 
(PART  2-PAR.  11.06} 


SUPPLY  VERTICAL  RUN  ON  ""^V'" 

PINS  AND  INSULATORS  ---- 

(PART  2  -PAR.  ti.05  (a))          ~~^^ 

NOTES:  """* 

1.  DRAWINGS   ARE  ILLUSTRATIVE  ONLY.   RE- 
QUIRED SEPARATIONS  MAY  NECESSITATE 
LOCATING  ATTACHMENTS  AT  OTHER  THAN 
THE  SAME  LEVEL 

2.  45"  WHERE   PRACTICABLE,  BUT  IN  NO  CASE 
SHALL  VERTICAL  RUNS  HAVE  A  CLEARANCE 
OF  LESS  THAN   2  INCHES  FROM  THE  NEAR- 
EST METAL  PART  OF  THE  EQUIPMENT  OF 
ANOTHER  PARTY    (PART  2-PAR,  11.02) 

PREFERENCES,  SUCH  AS  (PART2-PAR.II.08> 
REFER  TO  THE  TEXT  IN  A  REPORT  OF  THE 
JOINT  COMMITTEE  ON  PLANT  COORDINA- 
TION OF  THE  EDISON  ELECTRIC  INSTITUTE 
AND  THE  BELL  TELEPHONE  SYSTEM,  ENTI- 
TLED «  JOINT  POLE  PRACTICES  FOR  SUPPLY 
AND  COMMUNICATION  CIRCUITS."  ISSUED 
OCTOBER  29,  1945  STREET  SIDE 

FIG.  3.     Location  of  Vertical  Runs  (Joint  Pole  Practices  for  Supply  and  Communication  Circuits, 

Oct.  29,  1945) 

The  requirements  for  normal  and  special  joint  use  conform  to  Code  Grade  C  and  B 
construction,  respectively,  applying  mainly  to  supporting  structures,  supply  conductors, 
and  clearances. 

JOINT-USE  &TJRAL  LINE  CONSTRUCTION  at  the  higher  voltages  (above  5000 
volts)  for  power  and  telephone  services,  though  recognized  for  some  time  as  the  best 
engineering  solution  for  many  situations  of  power  and  telephone  lines  along  the  same  route 
and  though  employed  in  various  specific  cases,  has  presented  a  number  of  problems  in  co- 
ordination. In  particular,  the  longer  spans  and  higher  voltages  for  the  power  circuits  and 
the  increased  noise  induction  from  the  longer  exposures  than  are  normally  encountered  in 
joint-use  urban  construction  are  main  factors.  The  rapid  growth,  within  about  the  last 
decade,  of  rural  electrification  has  greatly  emphasized  the  necessity  for  a  study  and  solu- 
tion of  these  problems,  to  permit  a  more  general  application  of  rural  j  oint-use  construction. 

For  the  purposes  of  study  of  safe  and  economical  rural  joint-use  construction  at  the 
higher  voltages,  several  projects  of  this  type  were  completed  and  placed  in  service  before 
1947  in  Alabama,  in  the  light-loading  district,  and  in  Minnesota  and  South  Dakota,  both 
in  the  heavy-loading  district. 

The  NES  Code  (Fifth  Edition  of  Part  2)  provides  that,  if  the  supply  circuit  will  be 
promptly  de-energized  in  the  event  of  accidental  contact  with  the  telephone  plant,  and  the 
resulting  voltages  on  the  telephone  plant  from  such  a  contact  will  be  within  the  operating 
capabilities  of  the  telephone  protective  equipment,  Grade  C  construction  may  be  employed 
in  joint  use.  This  provision  of  the  Code  has  an  important  bearing  on  the  problems  involved 
in  long-span  rural  joint  use. 

For  long-span  construction,  high-strength  line  wires  are  necessary;  they  are  generally 
of  stranded  copper  or  aluminum  with  one  or  more  strands  of  steel  for  power  circuits,  and 
of  high-strength  solid  steel  or  copper-covered  steel  for  telephone  circuits.  The  minimum. 
size  of  power  wire  for  Grade  C  construction  under  the  Code  is  No.  8  AWG  medium-hard 
drawn  copper. 

With  high-strength  line  wires  for  both  power  and  telephone  conductors,  it  was  assumed 
that  their  sag  characteristics  would  be  sufficiently  alike  to  prevent  contacts  in  the  span 
under  ice  or  wind  loadings  or  temperature  changes,  with  reasonable  minimum  separations 


10-72 


TRANSMISSION  CIRCUITS 


of  the  two  classes  of  conductors  at  the  poles.  It  was  also  assumed  that  a  very  large  per- 
centage of  rural  power  circuits  would  consist  of  one  phase  wire  on  a  pin  at  the  top  of  the 
pole  and  a  multigrounded  neutral  below  it  on  a  pin  or  secondary  rack,  or  that  the  primary 
circuit  would  consist  of  two  or  more  primary  wires  on  a  cross-arm  at  the  top  of  the  pole 
and  that  the  multigrounded  neutral,  if  present,  would  be  located  on  the  same  cross-arm 
or  below  it. 

Where  joint  rural  lines  cross  fields  or  other  property,  which  is  or  is  likely  to  be  traversed 
by  loaded  vehicles  or  farm  machinery,  adequate  wire  clearances  should  be  provided. 


SUPPLY  CROSS  ARMS;;' 


COMMUNICATION 

CABLE 


GUYS  (NOT  PARALLEL  TO  LINE)  SHALL  CLEAR   SUPPLY 

LINE  CONDUCTORS  ATTACHED  TO  THE  SAME   POLE 
BY  AT  LEAST  6  INCHES   PLUS   QA  INCH   FOR  EACH 
fOOO  VOLTS   IN  EXCESS  OF    8700  VOLTS 
(PART2-PAR.16U33) 


__7    STRAIN   INSULATORS 
'-\RT  2 -PAR.  17.03  (a) (3)) 


GUYS  SHALL  CLEAR  COMMUNICATION  CABLES  OR  LINE 
CONDUCTORS  BT  AT  LEAST  8  INCHES  WHERE  PRACTI- 
CABLE AND  W  NO  CASE  LESS  THAN  3  INCHES 
(PART  e-FAR.  16.03) 


SUPPLY  CROSSARM 
V 


COMMUNI- 
CATION 
CROSSARM 


THIS  INSULATOR  MAY  BE  OMITTED  IF  THE 
POINT  OF  ATTACHMENT  TO  THE  STUB  IS 
MORE  THAN  8  FEET  ABOVE  GROUND 
(PART  2- PAR.  17.03  OKI)} 


TWS  INSULATOR  MAY  BE  OMITTED  }F  SUY  IS 
NOT  CARRIED  OVER,  OR  UNDER  OVERHEAD  SUPPLY 
CONDUCTORS  OF  MORE  THAN    3OO  VOLTS  TO  GROUND 
{OTHER  THAN  THOSE  ON   THE  GUYED  POLE) 


NOTE'  REFERENCES,  SUCH  AS  (PART  2- PAR.  17 03)  REFER  TO  THE 
TEXT  IN  A  REPORT  OF  THE  JOFNT  COMMITTEE  ON  PLANT 
COORDINATION  OF  THE  EDISON  ELECTRIC  INSTITUTE  AND 
THE  BELL  TELEPHONE  SYSTEM,  ENTITLED  » JOINT  POLE 
PRACTICES  FOR  SUPPLY  AND  COMMUNICATION  CIRCUITS," 
ISSUED  OCTOBER  2?, 1945 


'^^^^ 

Use  of  Strain  Insulators  in  Ungrounded  Guys  (Joint  Pole  Practices  for  Supply  and  Communi- 
cation Circuits,  Oct.  29,  1945) 

Span  lengths  in  the  Alabama  installations  of  rural  joint  use  average  about  400  ft  with 
maximum  spans  of  600  to  800  ft.  In  Minnesota  and  South  Dakota,  the  loading  conditions 
being  more  severe,  the  average  and  maximum  span  lengths  are  about  320  ft  and  400  ft 
respectively. 

For  noise  induction  reasons,  the  separations  between  the  power  and  telephone  circuits 
should  be  kept  as  uniform  as  practicable  without  incurring  undue  costs  or  construction 
difficulties.  (See  article  31.) 

Electrical  protection,  to  meet  requirements,  should  provide:  (a)  fuses,  circuit-breakers, 
or  other  devices,  which  will  promptly  and  reliably  de-energize  the  power  circuit,  if  a  ground 
fault  of  relatively  low  impedance  occurs;  (6)  protective  gaps,  to  be  connected  between  the 
telephone  circuit  and  the  common  neutral  (or  other  low-resistance  ground)  at  such  intervals 
that  the  impedance  to  ground  of  the  telephone  plant,  in  case  of  a  power  wire  contact,  will 
be  low  enough,  to  permit  proper  operation  of  the  power  protective  devices. 

As  experience  indicates  that  the  frequency  of  contacts  is  very  low,  the  power  protective 
arrangements  are  usually  more  than  adequate  to  meet  the  above  requirements.  The  tele- 
phone circuits  in  the  joint  projects,  referred  to  above,  employ  protectors  having  a  3-kv 
(mis)  breakdown  to  ground,  placed  at  about  Va-mile  intervals,  thus  providing  a  low-im- 
pedance path  to  ground.  The  current-carrying  capacity  of  the  protector  must  be  sufficient 
$&  meet  tike  discharge  which  may  result  from  a  contact. 


INDUCTIVE   COORDINATION  10-73 

As  a  means  of  limiting  induced  60-cycle  open-circuit  voltages  (under  normal  operation) 
on  telephone  circuits,  which  are  occasionally  disconnected  from  the  central  office  equip- 
ment and  entrance  cables,  a  drainage  device,  consisting  of  a  0.25-juf  condenser  in  series 
with  a  10,000-ohm  resistor  from  each  wire  to  ground,  may  be  used. 

Although  experience  at  present  is  limited,  the  feasibility  of  higher-voltage  long-span 
joint  use  in  rural  areas  has  been  established,  and  economic  studies  in  progress  indicate 
that  worth-while  economies  are  possible  for  most  new  extensions  and  sometimes  even 
where  power  lines  have  already  been  built. 

HIGHER-VOLTAGE  CROSSINGS  (involving  supply  lines  of  more  than  5000  volts 
between  wires  or  2900  volts  to  neutral  or  ground)  are  given  special  consideration  to  min- 
imize possible  contacts  between  supply  and  communication  circuits.  The  Code  require- 
ments for  spacings,  clearances,  and  strength  of  construction  are  considered  to  be  minimum 
and  represent  the  generally  accepted  practices  (except  as  modified  by  more  exacting  state 
or  local  regulations)  throughout  the  United  States. 

RURAL  POWER  LINE  CARRIER  TELEPHONE  SYSTEMS,  where  applicable,  are 
being  placed  in  operation  throughout  the  country  as  a  means  of  providing  telephone  service 
to  rural  subscribers  who  are  accessible  from  rural  power-line  systems  but  not  readily  so 
from  rural  telephone  lines.  Although  these  carrier  telephone  systems  are  still  in  their 
early  trial  periods,  it  appears  that  they  will  eventually  be  economically  useful  in  furnishing 
telephone  service  to  distant  farms  and  ranches  which  would  otherwise  require  costly 
telephone-line  construction  to  reach.  Joint  practices  and  agreements  covering  this  type 
of  operation  are  in  the  formative  stage. 

A  more  detailed  description  of  this  carrier  system  and  its  associated  installation  and 
operating  features  is  given  in  Section  17. 

30.  INDUCTIVE  COORDINATION 

Inductive  coordination,  as  it  applies  to  supply  (electric)  and  communication  companies, 
embraces  the  principles  and  practices,  agreed  upon  by  these  companies,  for  the  design, 
location,  construction,  operation,  and  maintenance  of  their  respective  systems,  located  in 
the  same  general  territory,  in  such  manner  as  to  prevent  interference  with  the  furnishing 
of  either  the  electric  or  communication  service. 

Since  much  of  the  general  public  throughout  the  country  receives  both  electric  and  com- 
munication service  by  means  of  overhead  construction,  it  is  inevitable  that  many  situations 
of  proximity  between  the  two  services  will  be  created  under  a  wide  variety  of  construction 
and  operating  conditions.  Also,  telephone  circuits  usually  transmit  relatively  low  amounts 
of  electrical  speech  power  (varying  from  less  than  1  to  a  few  milliwatts) ,  whereas  electric 
transmission  and  distribution  circuits  transmit  power  ranging  from  a  few  to  hundreds  and 
thousands  of  kilowatts.  Owing  to  the  much  greater  amounts  of  power  carried  by  these 
circuits,  not  only  at  the  fundamental  but  also  in  many  cases  at  harmonic  frequencies  within 
the  voice-frequency  range,  inductive  effects  from  power  circuits  may,  under  certain  condi- 
tions of  proximity  and  operation,  severely  affect  com  muni  cat  ion  service,  if  proper  control 
is  not  provided. 

The  control  of  inductive  interference  in  communication  circuits,  to  permit  giving  a 
satisfactory  service,  is  accomplished  through  the  application  of  general  coordinated 
methods,  or,  when  required,  by  applying  specific  coordinated  methods,  as  set  forth  in  the 
Principles  and  Practices,  discussed  in  article  28. 

There  are  two  general  classifications  of  inductive  interference,  namely:  (a)  noise  fre- 
quency (induction  from  power  harmonics  within  the  voice-frequency  range);  (&)  low- 
frequency  (induction  from  the  power  fundamental  frequency,  usually  60  cycles,  during 
abnormal  power  circuit  conditions). 

Inductive  effects  between  power  and  communication  circuits  arise  from  the  fact  that 
power  wires,  transmitting  relatively  large  a-c  voltages  and  currents,  establish  strong  elec- 
tric and  magnetic  fields  in  their  vicinity,  which,  owing  to  their  varying  character,  set  up 
in  nearby  communication  wires  alternately  increasing  and  decreasing  electric  voltages  and 
currents  of  the  same  frequencies  as  those  in  the  power  wires.  Induction  from  communica- 
tion to  power  wires  would  have  no  noticeable  effect  in  any  event  on  power  service,  because 
of  the  small  amount  of  power  carried  by  the  communication  wires  and  the  nature  of  the 
power  service. 

It  is  often  desirable  to  consider  effects  of  magnetic  and  electric  induction  separately, 
particularly  in  the  technical  analyses  of  specific  problems.  This  is  not  only  because  the 
physical  processes  and  the  effects  of  voltage  and  current  induction  are  quite  different  but 
also  because  the  power-circuit  voltages  and  currents  are  often  affected  differently  by 


10-74 


TRANSMISSION  CIRCUITS 


•esenting 


FIG.  5.     Diagram  Showing  Couplings  between  Power 
and  Telephone  Wires  for  Electric  Induction 


changes  in  conditions.    Electric  induction  is  due  to  the  voltages  on  the  power  line;  mag- 
netic induction  is  due  to  currents  on  the  power  line. 

Theoretically,  electric  and  magnetic  induction  are  produced  as  described  briefly  below. 

A  simple  method  of  visualizing  electric  induction  is  by  means  of  the  capacitances  in- 

volved with  a  single  power  wire  and  a  single  telephone  wire,  as  shown  in  Fig.  5.    Neglecting 

the  impedances  outside  the  exposure  (shown  dotted  in  Fig.  5),  the  voltage  of  the  power 

wire  to  ground  (Ep)  divides  over  the  capacitances  CTP  and  CTO  in  proportion   to  their 

impedances  (that  is,  in  inverse  ratio  to 
their  capacitances)  .  The  induced  voltage 
(ET)  on  the  telephone  wire  may  therefore 
be  expressed  mathematically  as: 

^-tsrns;*      (1) 

Where  there  are  numerous  power  and 
telephone  wires,  capacitances  exist  be- 
tween every  possible  combination  of  wires, 
and  of  wires  and  ground,  resulting  in  a 
complicated  network,  but  the  principles 
involved  are  the  same  as  in  the  simple  case 
discussed  above. 

The  potential  of  the  telephone  wire 

to  *»  the  sam/  ^  a!°nS  ^length. 
the  wire  is  perfectly  insulated  from 
ground,  extends  only  through  the  length 
of  the  exposure,  and  has  no  equipment  on  it,  this  potential  is  independent  of  the  length 
of"  the  exposure  (the  condition  shown  in  Fig.  5  if  the  impedances  to  ground  are  neglected). 
This  is  true  because,  whereas  all  the  capacitances  in  the  above  equation  are  proportional 
to  exposure'  length,  the  ratio  CTP/(CTQ  +  CTP)  is  independent  of  length.  However,  in 
practice,  the  circuits  usually  extend  beyond  the  exposure  and  have  equipment  connected 
between  them  and  ground,  so  that  there  are  impedances  to  ground  outside  the  exposure 
through  which  longitudinal  current  will  flow.  Since  the  impedance  of  CTP  controls  the 
total  longitudinal  current,  this  current  will  be  practically  independent  of  the  telephone- 
circuit  impedances  to  ground  and  will  be  proportional  to  exposure  length.  It  will  also 
be  proportional  to  the  frequency  of  the  harmonics  in  the  inducing  voltage.  Hence,  for 
given  telephone-circuit  impedance  conditions  (outside  the  exposure),  the  voltage  to  ground 
will  be  proportional  to  exposure  length  and  to  the  frequency  of  the  inducing  voltage  in  a 
uniform  and  electrically  short  exposure. 

In  magnetic  induction,  the  current  in  the  power  wire  sets  up  a  magnetic  field  which 
alternates  at  the  frequency  of  the  current.  If  a  communication  wire  is  located  in  this  field, 
a  wltage  is  induced  along  it  which  is  pro- 
portional to  the  rate  of  change  of  the  mag- 
netic flux.  This  phenomenon  is  illustrated 
in  Fig.  6.  The  voltage  between  the  tele- 
phone "  circuit  and  ground  varies  from 
point  to  point  along  the  circuit  and  de- 
pends on  the  distribution  of  the  imped- 
ances to  ground  as  well  as  on  the  distribu- 
tion of  the  induced  voltage.  Also,  since 
the  voltage  acts  along  the  circuit  and  the 
part  induced  in  each  short  length  adds 
directly  to  those  in  all  other  short  lengths, 
the  total  induced  voltage  is  directly  pro- 
portional to  the  exposure  length  in  a  uniform  and  electrically  short  exposure.  Further, 
as  the  rate  of  change  of  magnetic  fiux  is  proportional  to  frequency,  the  induced  voltage 
will  be  J>roportional  to  the  frequency  of  the  inducing  current. 

In  the  foregoing,  the  factors  discussed  apply  to  both  noise  and  low-frequency  induction. 
However,  these  two  general  types  of  problems  are  discussed  separately. 


Magnetic  Flux 


FIG.  6.    Diagram  Showing  Coupling  between  Power 
and  Telephone  Wires  for  Magnetic  Induction 


31.  NOISE  FREQUENCY  INDUCTION 

Noise  frequency  induction,  in  communication  circuits,  for  a  given  situation  of  proximity, 
depends  upon: 

(a)  Inductive  influence  of  the  power  circuits,  as  determined  by  their  electrical  char- 
acteristics. 


NOISE  FREQUENCY  INDUCTION 


10-75 


(b)  Inductive  coupling  between  the  power  and  communication  circuits,  as  determined 
by  their  physical  relation. 

(c)  Inductive  susceptiveness  of  the  communication  circuits,  as  determined  by  their 
electrical  characteristics. 

INDUCTIVE  INFLUENCE.  Two  characteristics  of  a  power  system  of  primary  im- 
portance in  determining  its  inductive  influence  are  wave  shape  and  balance. 

The  wave  shape  of  the  voltage  or  current  on  a  power  line  is  a  function  of  the  magnitudes 
and  frequencies  of  the  harmonics,  which  may  induce  voltages  of  frequencies  within  the 
range  ordinarily  used  in  telephone  circuits.  Induced  voltages  at  such  frequencies  have 
much  greater  interfering  effects  than  the  voltage  induced  at  the  fundamental  frequency. 

The  approximate  relative  interfering  effects  of  telephone  line  voltages  and  currents  is 
shown  in  Fig.  7  for  certain  subscriber  sets  and  instruments  employed  in  the  Bell  System. 


o 

5 

—•—LINE  WEIGHTING  ADOPTED  1935 
WHEN  TELEPHONE  SETS  EMPLOYED 
\44  OR    557  RECEIVERS. 
1941  LINE  WEIGHTING  FOR  F1A-101 
SETS    (USING  HAI  RECEIVERS) 

1 

\ 

>7 

\ 

*^. 

j  10 

P 

|25 
£30 

S3* 
40 
-45 

/ 

/ 

\ 

"**" 

\ 

/ 

/ 

^X 

^ 

/ 

/ 

\\ 

/ 

/ 

M 

/ 

/ 

V\ 

/ 

1 

/ 

1  ] 

15O2OO    4OO  6OO  tOOO    2OOO   4OOO 
,   '  «}EQUENCY  W  CYCLES  PER  SECOND 

FIG.  7.  Line  Weightings  for  Tele- 
phone Message  Circuit  Noise  (Rela- 
tive Interfering  Effects  of  Telephone 
Line  Voltages  or  Currents)  (Report 
45,  J.C.P.C.) 


0 
5 

w  K> 

§  is 

*» 

S25 
g30 

C35 

•40 
45 

—  —  RECEIVER  WEIGHTING  ADOPTED 
ttJ  1935   (J44  RECEIVER)  * 

—  J941  RECEIVER  WElGHTlNC 
(HA)  RECEIVER) 

^*~ 

/ 

t- 

•^\ 

/ 

^ 

/ 

Vs 

\ 

t 

X 

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s 

s 

\ 

N 

NV 

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t 

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1 

1   1 

ISO  2OO          4OO    6OO     100O        2OOO         AOOO 
FREQUENCY  IN  CYCLES  PER  SECOND 

*Also  applicable   to  557  receiver  for 
estimating  receiver  noise. 

FIG.  8.  Receiver  Weightings  for  Tele- 
phone Message  Circuit  Noise  (Relative 
Interfering  Effects  of  Telephone  Line 
Voltages  or  Currents  at  Receiver)  (Re- 
ports 32  and  45,  J.C.P.C.) 


The  solid  curve  shows  the  line  weighting  adopted  in  1941  for  the  present-day  anti-sidetone 
set  having  a  relatively  flat  response  over  the  useful  voice-frequency  range.  This  curve 
also  takes  into  account  (1)  the  possible  utilization,  in  future  telephone  receiving  elements, 
of  a  larger  part  of  the  wide-band  toll  transmission  attained  in  the  later-type  carrier  toll 
circuits,  (2)  a  more  uniform  response  within  the  transmission  band,  and  (3)  the  attenuation 
at  frequencies  near  3000  cycles  and  higher,  relative  to  that  at  1000  cycles,  caused  by  var- 
ious types  of  switching  trunks  used  in  toll  connections,  as  compared  to  the  distortionless 
trunks  used  in  the  tests  on  which  this  curve  is  based.  The  two  curves  cannot  be  compared 
on  an  absolute  basis,  since  the  acoustic  output  per  volt  input  at  1000  cycles  is  not  the  same 
for  the  different  sets. 

Receiver  weighting  curves  for  telephone  message  circuit  noise  are  shown  in  Fig.  8  for 
certain  instruments  employed  in  the  Bell  System.  These  curves  differ  from  the  line  weight- 
ing curves,  principally  as  a  result  of  taking  into  account  the  voltage  loss,  relative  to  that 
at  1000  cycles,  caused  by  the  trunk,  loop,  and  telephone  set,  between  the  toll  board  (at 
which  point  the  line  weightings  apply)  and  the  telephone  set  receiver. 

Each  harmonic  voltage  or  current  induced  in  a  telephone  circuit  by  a  paralleling  power 
circuit  will  individually  react  in  the  telephone  circuit,  over  which  a  conversation  may  be 
in  progress,  and  in  such  a  manner  that  the  listener  hears  the  combined  effect  of  all  the 
harmonics,  which  is  termed  noise.  Different  frequencies  in  this  noise  have  different 
interfering  effects,  depending  on  the  characteristics  of  the  telephone  circuits,  type  of 
receiver,  the  human  ear,  and  other  factors.  Relative  interfering  effects  (Fig.  8)  are  called 
noise  weightings  and  have  been  determined  by  extensive  tests. 

Since  the  effects  of  electric  and  magnetic  couplings  are  directly  proportional  to  frequency, 
the  relative  noise  influence  of  power-system  voltages  and  currents  is  proportional  to  the 
product  of  noise  weighting  and  frequency.  For  any  frequency,  this  product  times  a  con- 
stant gives  the  telephone  influence  factor,  TIF.  TIF  weightings  versus  frequency  are 
shown  in  Fig.  9  for  three  different  periods,  reflecting  changes  in  transmission-frequency 
characteristics  of  telephone  circuits  and  instruments  from  1918  to  1941.  The  1941  curve 


10-76 


TRANSMISSION  CIRCUITS 


has  not  yet  been  standardized.    It  will  be  noted  from  Fig.  9  that  the  TIF  values  at  60 
cycles  are  very  low. 

While  a  great  deal  of  inductive  coordination  work  makes  use  of  single  harmonic  fre- 
quency data,  there  are  cases  where  it  is  desirable  to  evaluate  the  overall  influence  of  a 
voltage  or  current  in  a  power  circuit  in  terms  of  a  single  measure.  Such  a  measure  may 
be  obtained  by  multiplying  the  magnitude  of  each  harmonic  present  (amperes  for  current 


14,000 


V 


1941  (TENTATIVE) 


12OO        16OO        2000        2400        2SOO 
FREQUENCY  IN  CYCLES   PER  SECOND 


3200         3600       4000 


FIG.  9.    TIF  Weightings  for  Periods  1918,  1935,  and  1941  (Courtesy  A.I.E.E.) 

and  Mlovolts  for  voltage)  by  its  TIF  weighting,  and  taking  the  rss  value  (square  root  of 
the  sum  of  the  squares)  of  these  products.  The  result  for  current  is  the  I  •  T  product,  and 
for  voltage  it  is  the  KV-T  product. 

These  products  can  be  measured  directly  with  a  suitable  noise-measuring  set  and  a 
current  or  voltage  TIF  coupler,  the  latter  having  a  transmission-frequency  characteristic 
directly  proportional  to  frequency. 

The  magnitudes  and  frequencies  of  the  harmonic  currents  and  voltages  on  a  power  line 
depend  on  the  characteristics  of  the  apparatus  and  associated  equipment  and  on  the 
impedance  of  the  supply  line  at  each  of  the  harmonic  frequencies.  The  wave  shape  at 
various  points  on  the  power  system  depends  on  the  way  in  which  the  various  harmonic 
currents  and  voltages  are  propagated  over  the  system.  Since  power  systems  are  usually 
very  complex  electrically,  the  propagation  effects  may  vary  greatly  for  different  frequencies 
and  for  different  systems. 

It  is  impracticable  to  construct  rotating  electrical  machinery  or  power  transformers 
entirely  free  from  harmonics,  although  marked  progress  has  been  made  in  this  respect. 
Also,  it  is  inherent  in  the  operation  of  rectifying  devices  (and  some  other  types  of  devices 
where  the  current  is  not  directly  proportional  to  voltage)  that  harmonics  are  produced. 
Generally  speaking,  the  factors  affecting  the  production  of  harmonics  in  these  general 
classes  of  apparatus  are  as  follows:  ~~" 

Motors  and  Generators.  Harmonics  are  affected  by  the  distribution  of  air-gap  flux, 
variations  in  the  air-gap  flux  due  to  the  slots  in  the  rotors  and  stators,  the  distribution 
of  the  windings  on  the  armature,  and,  in  multiphase  machines,  connections  of  the  windings. 

Transformers.  The  degree  of  saturation  of  the  iron  in  the  core  affects  the  harmonics 
materially.  In  polyphase  transformer  banks,  the  connection  of  the  transformers  in  the 
bank  affects  some  of  the  harmonics,  particularly  the  triple.  In  a  3-phase  transformer  the 
arrangement  of  the  core 
also  affects  the  triple 
harmonics. 

Rectifiers.  Owing  to 
the  fiat-top  wave  shape 
of  rectifier  anode  cur- 
rents, the  a-c  line  cur- 
rent, taken  by  the  recti- 
fier, has  a  step-type 
wave  dbape,  as  shown  in 


12- PHASE   RECTIFIER 


FIG.   10.     A-c  Line   Currents  Taken   by  6    and   12   Phase   Rectifiers 


NOISE  FREQUENCY  INDUCTION 


10-77 


Fig.  10  for  a  6-  and  12-phase  rectifier.  This  wave  shape  results  in  the  production  of 
harmonic  currents  and  voltages  in  the  a-c  line.  Table  1  shows  the  order  of  these  harmonics 
(multiples  of  the  fundamental  frequency)  for  various  multiphase  rectifiers  with  balanced 
operation,  and  also  the  reduction,  with  increase  in  phases,  of  the  number  of  harmonics 
which  are  of  importance. 

The  magnitude  of  any  harmonic  which  is  present  with  any  particular  number  of  phases 
is  the  same  as  for  a  6-phase  rectifier,  assuming  all  other  conditions  to  be  the  same.    Thus, 
the  magnitude   of   the   twenty- 
Table  1. 


third  harmonic  is  the  same  for 
the  6-,  12-,  and  24-phase  recti- 
fiers. Also,  the  magnitudes  of 
the  harmonic  currents  bear  a 
definite  relation  to  the  rms  value 
of  the  total  rectifier  current  and 
decrease  in  value  with  increase  in 
frequency.  For  a  given  kilovolt- 
ampere  input  to  a  rectifier,  the 
higher  the  a-c  line  voltage  the 
lower  is  the  a-c  line  current,  with 
a  corresponding  reduction  in  the 
magnitudes  of  the  harmonic  cur- 
rents and  usually  in  the  harmonic 
voltages  resulting  from  them. 
Furthermore,  there  are  generally, 
under  modern  practices,  fewer 
long  closely  coupled  exposures 
with  the  high-voltage  transmis- 
sion circuits  than  with  the  lower- 
voltage  power  distribution  net- 
works. 

Phase  control,  employed  on 
rectifiers  for  reducing  the  d-c  out- 
put voltage  below  that  obtained 
without  phase  control,  is  accom- 
plished by  retarding  the  firing 
point  of  the  anodes  in  the  alter- 


Harmonics  Arising  in  Rectifiers 


Orders  of  Harmonics  in  Line  with 
Balanced  Operation 

Corresponding 
Harmonic 

Rectifier  Phases 

Frequencies 
on  60-cyele 

6 

12 

18 

24 

36 

48 

System 

5 

300 

7 

420 

11 

11 

660 

13 

13 

780 

17 

17 

1020 

19 

19 

H40 

23 

23 

23 

1380 

25 

25 

25 

1500 

29 

1740 

31 

1860 

35 

35 

35 

35 

2100 

37 

37 

37 

37 

2220 

41 

2460 

43 

2580 

47 

47 

47 

47 

2820 

49 

49 

49 

49 

2940 

53 

53 

3180 

55 

55 

3300 

59 

59 

3540 

61 

61 

3660 

Note:  Higher  harmonics  are  also  present  for  all  types  listed. 


nating-voltage  cycle,  through  grid  or  firing  control.  The  power  factor  at  the  a-c  line 
terminal  of  the  rectifier  transformer  is  lowered.  The  magnitude  of  the  harmonic  com- 
ponents increases  in  the  a-c  line  current  for  a  given  kilowatt  output  of  the  rectifier. 
Also,  with  phase  control,  the  anode  currents  have  a  steeper  wave  front  at  the  beginning 
and  end  of  the  anode  firing  period,  during  commutation  between  successive  anodes,  result- 
ing in  harmonics  of  higher  magnitude.  Phase  control  should,  therefore,  be  limited  to  actual 
requirements,  particularly  at  full  load  and  overload  ratings  of  rectifiers,  in  order  to  limit 
the  possible  inductive  interference. 

Balanced  and  Unbalanced  Currents.  In  a  multiphase,  balanced  power  circuit  the 
voltages  between  the  several  phase  conductors  and  between  the  phase  conductors  and 
ground,  and  also  the  several  line  currents,  are  vectorially  equal  to  zero. 

When  the  currents  or  voltages  do  not  vectorially  equal  zero,  they  contain  a  set  of  single- 
phase  components,  all  in  the  same  phase  relation,  which  are  termed  residual  components. 
Any  system  of  voltages  or  currents  can  be  resolved  into  its  balanced  and  residual  com- 
ponents, and  the  effects  of  each  can  be  analyzed  separately.  The  balanced  components  are 
confined  wholly  to  the  phase  conductors;  the  residual  components  act  in  a  path  consisting 
of  the  phase  conductors  and  an  external  return  as,  for  instance,  a  metallic  neutral  or 
through  the  earth.  Since  the  coupling  for  the  residual  components  is  usually  much  larger 
than  for  the  balanced  components,  the  former  are  usually  of  greater  importance  in  coordi- 
nation problems. 

Single-phase  branches  on  three-phase  distribution  systems  are,  of  themselves,  inherently 
unbalanced.  On  grounded  neutral  systems  the  residual  voltage  on  a  single-phase  branch 
is  practically  equal  to  the  phase-to-neutral  voltage.  On  isolated  neutral  systems  the 
residual  voltage  on  a  single-phase  branch  depends  on  the  particular  system  layout.  The 
single-phase  branches  also  introduce  residual  currents  and  voltages  on  the  3-phase 
system. 

With  the  present  methods  of  analyzing  noise  induction  problems,  the  balanced  and 
residual  currents  and  voltages  are  usually  considered  separately.  In  the  general  case  of 
exposures  of  overhead  lines  of  the  multigrounded  neutral  type  to  subscribers'  cable  cir- 


10-78 


TRANSMISSION  CIRCUITS 


Table  2 


cults,  a  knowledge  of  the  residual  currents  is  sufficient,  the  effect  of  the  balanced  currents 

being  relatively  unimportant. 

INDUCTIVE  COUPLING.    The  coupling  between  power  and  communication  circuits 

is  determined  by  the  degree  of  their  proximity,  but  it  may  be  greatly  modified  by  the  bal- 
ance of  the  two  classes  of  cir- 
cuits to  each  other  and  by  the 
proximity  of  grounded  linear 
circuits  or  metallic  objects. 

In  determining  coupling,  it 
is  desirable  to  differentiate 
between  the  effects  of  the  bal- 
anced and  residual  compo- 
nents in  the  power  circuit,  be- 
tween the  effects  of  voltages 
and  those  of  currents,  and,  on 
the  telephone  line,  between 
induced  voltage  which  acts 
directly  in  the  metallic  cir- 
cuit, termed  metallic-circuit 
induction,  and  that  which  acts 
in  the  circuit  composed  of  the 
wires  in  parallel  with  ground 
return,  termed  longitudinal- 
circuit-induction. 

Types  of  Induction.  Eight 
components  of  power  induc- 
tion, shown  in  Table  2,  need 


Types  of  Induction 

Transpositions  Tending 

to  Reduce 
Induction  (I  =  Telephone 

I 

f  =  rower 

A. 

Metallic-circuit  (direct) 

I.  From  balanced  currents 

T 

2.  From  balanced  voltages 

T 

3.  From  residual  currents 

T 

4.  From  residual  voltages 

T 

B. 

Longitudinal-circuit  (indirect 

metallic-circuit)  * 

5.  From  balanced  currents 

P 

6.  From  balanced  voltages 

P 

7.  From  residual  currents 

P 

)  Only  if  residuals  are 

8.  From  residual  voltages 

P 

)      thereby  reduced. 

*  This  component  of  induction  can  result  in  noise  in  the  metallic 
circuit,  because  of  the  reaction  of  such  longitudinal  induction  upon 
self-unbalances  (high-resistance  joints  or  leakage)  or  mutual  induct- 
ance or  capacitance  unbalances  to  other  wires  on  the  line  or  to 
ground. 


to  be  considered  in  noise  induction  problems.    These  components  vary  in  their  importance, 
as  noise  factors,  for  different  situations  and  for  the  reasons  discussed  above. 

Transpositions  are  employed  in  open  wires  within  exposures  (parallels)  between  power 
and  communication  circuits  as  an  aid  in  neutralizing  induced  power-circuit  noise  in  the 
latter  circuits.  Transpositions  are  required  generally  in  open- wire  communication  circuits, 
whether  or  not  power-circuit  exposures  exist,  to  limit  intercircuit  cross-talk. 


Physical  or  Side  Circuit 


X 


Types  of  Phantom  Transposftions 


Typel 


Type  2 


Type  3 


Telephone  Circuit  Transpositions 


Type  4 


Single  Pha 


Three  Phase 


Power  Circuit  Transpositions 
FIG.  II.     Diagrams  Showing  Wire  Arrangements  at  Transposition  Points 


NOISE  FREQUENCY  INDUCTION 


10-79 


Transpositions  are  made  By  interchanging  in  a  uniform  manner  the  positions  of  the 
wires  comprising  a  circuit,  so  that  each  wire  of  the  circuit  occupies  all  the  pin  positions 
occupied  by  the  circuit,  for  distances  (usually  equal)  as  determined  by  the  transposition 
design.  Figure  11  shows  the  changes  employed  in  the  position  of  the  wires  (both  telephone 
and  power)  at  transposition  points.  Telephone  transpositions  may  be  of  the  physical  or 
side  circuit  types  involving  two  wires  or  of  any  one  of  four  types  of  phantom  transpositions 
involving  four  wires.  Power  transpositions  may  be  of  the  single-phase  type  involving 
two  wires  or  of  the  3-phase  type,  involving  three  wires. 

Identical  3-phase  power  transpositions,  when  placed  at  the  l/s  and  2/3  points  in  a  given 
uniform  section  of  power  line,  establish  a  power  circuit  barrel,  since  each  wire  is  rotated 
120°  in  phase  position  and  in  the  same  direction  of  rotation  at  each  transposition.  Single- 
phase  power  transpositions  rotate  the  wires  ISO0  in  phase  position.  Likewise,  for 
metallic-circuit  induction,  telephone  transpositions  change  the  phase  of  the  induction  by 
180°. 

Figure  12  shows  a  simple  arrangement  of  telephone  and  power  transpositions  within  a 
unit  exposure  length,  Z,,  containing  one  power  barrel.  In  this  arrangement,  which  is 


Y                           Y 

A                            A 

.      L_     ,  ^                         L                          j^ 

1.                 fc 

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3                            * 

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Power 


Telephone 


a  *>•< NEUTRAL  POINTS™*-  C  O 

Y    »   TRANSPOSITION  OF  ALL  PHASE-CONDUCTORS 
tl  OF  A  .THREE-  PHASE  CIRCUIT 

FIG.  12.     Simple  Balanced  Arrangement  of  Telephone  and  Power  Transpositions  in  a  Unit  Exposure 
Length,  L  (Courtesy  Bell  System) 

commonly  employed,  the  3-phase  power  transpositions  are  located  at  neutral  points  with 
respect  to  the  telephone  transpositions,  and  the  telephone  transpositions  are  so  arranged 
as  to  reduce  the  direct  metallic-circuit  induction  (from  both  balanced  and  residual  com- 
ponents) within  each  section  of  exposure  between  power  transpositions,  as  well  as  to  limit 
inter-cross-induction . 

Telephone  circuit  transpositions  tend  to 

(a)  Reduce  intercircuit  mutual  effects,  known  as  cross-induction  or  cross-talk. 

(6)  Reduce  direct  metallic-circuit  induction  from  both  balanced  and  residual  com- 
ponents of  power  circuits,  within  exposures,  especially  when  coordinated. 

(c)  Balance  the  two  sides  of  the  telephone  circuit  with  respect  to  earth  and  with  respect 
to  all  other  wires  on  the  telephone  line,  considered  as  one  longitudinal  circuit  (Sigma) . 

These  transpositions,  by  themselves,  are  not  effective  in  reducing  the  longitudinal-circuit 
induction  from  either  the  balanced  or  residual  components  of  the  power-circuit  voltages 
and  currents.  However,  within  an  inductive  exposure,  such  transpositions  tend  in  some 
measure  to  equalize  these  inductive  effects  by  exposing  each  conductor  of  the  telephone 
circuit  equally  to  the  power-circuit  influences. 

For  telephone-circuit  transpositions  to  be  reasonably  effective  in  reducing  metallic- 
circuit  induction,  the  relation  between  the  power  and  telephone  circuits  within  each  co- 
ordinated section  of  exposure  (with  respect  to  each  other,  to  ground,  and  to  other  circuits 
present)  must  be  substantially  uniform.  Thus,  points  of  discontinuity  within  an  exposure, 
such  as  sharp  changes  in  separation,  crossings,  or  changes  in  power-circuit  configuration, 
must  be  considered  in  inductive  coordination  work. 

Power-circuit  transpositions,  within  the  exposure,  tend  to  reduce  the  longitudinal- 
circuit  induction  from  balanced  voltages  and  currents  and,  as  a  result,  that  component  of 
metallic-circuit  noise  arising  mainly  outside  the  exposure  due  to  the  action  of  the  longi- 
tudinal-circuit induction  (from  balanced  components)  upon  any  unbalances  affecting  the 
telephone  circuits. 

Notes.  1.  Power  transpositions,  in  the  usual  case,  do  not  appreciably  affect  the  direct  metallic- 
circuit  induction. 

2.  That  part  of  the  direct  metallic-circuit  induction  which  results  from  residual  voltages  and  currents 
is  not  affected  by  power-circuit  transpositions  except  in  so  far  as  such  transpositions  may  reduce  the 
power-circuit  residuals 


10-80 


TRANSMISSION  CIRCUITS 


INDUCTIVE  SUSCEPTIVENESS.  The  degree  to  which  telephone  transmission  is 
adversely  affected  by  noise-frequency  induction  depends  not  only  upon  the  magnitudes 
of  the  induced  noise  voltages,  as  determined  by  influence  and  coupling  factors,  but  also 
upon  the  susceptiveness  factors  of  the  telephone  system.  These  include  the  manner  in 
which  the  induced  voltages  and  currents  are  propagated  to  the  circuit  terminals,  together 
with  the  reactions  of  the  circuit  unbalances  (thus  relating  the  current  in  the  terminal 
apparatus  to  the  induced  voltages),  the  sensitivity  of  the  receiving  apparatus,  and  the 
operating  power  level  of  the  telephone  circuits. 

Propagation  Effects  and  Balance.  Important  differences  exist  with  respect  to  propaga- 
tion effects  and  balance  between  open-wire  and  cable  circuits  and  between  toll  and  ex- 
change systems. 

As  pointed  out  in  the  discussion  of  coupling,  only  the  magnetically  induced  longitudinal 
voltages  and  currents  are  important,  under  the  conditions  usually  encountered,  in  produc- 
ing noise  in  telephone  cable  circuits.  Because  of  the  negligible  effects  of  electric  induction 
and  direct  metallic-circuit  induction  and  because  of  the  important  shielding  effects  exerted 
by  the  cable  sheath  and  the  various  telephone  circuits  on  each  other,  telephone  cable 
circuits  are  much  less  susceptive  than  open-wire  circuits. 

In  open-wire  telephone  systems,  consideration  must  be  given  both  to  electric  and  mag- 
netic induction  and  to  voltages  induced  directly  in  the  metallic  circuit  as  well  as  to  those 
induced  in  the  longitudinal  circuit.  In  a  line  composed  of  a  number  of  circuits,  the  currents 
set  up  in  any  one  circuit  depend  not  only  upon  the  voltage  induced  in  that  circuit  and  its 
impedance  but  also  upon  the  currents  and  voltages  which  are  set  up  in  the  other  telephone 
circuits  on  the  line.  It  is  not  possible,  therefore,  to  calculate  precisely  the  induced  currents 
merely  from  a  knowledge  of  the  magnitude  of  the  currents  and  voltages  on  the  power 
circuits  and  the  coupling  between  the  power  circuits  and  isolated  pairs  of  wires  on  the 
telephone  line,  considered  independently. 

Estimates  of  receiver  noise  for  a  particular  type  of  subscriber  set  and  receiver  connected 
to  a  given  line  circuit,  in  which  noise  currents  are  assumed  to  be  present,  may  be  made  as 
discussed  below. 

On  the  basis  of  the  definition  of  reference  noise,  in  absolute  terms,  as  an  electrical  power 
of  10~12  watt  at  a  frequency  of  1000  cycles  dissipated  in  a  receiver  (for  all  types  of  receivers) , 
Table  3  gives  the  currents  in  three  types  of  receivers  and  the  voltages  across  them  at  1000 

cycles  for  reference  noise  (0  db) 

Table  3  and  representative  values  of  the 

receiver  impedances. 

At  frequencies  other  than  1000 
cycles,  the  voltage  for  reference 
noise  is  the  voltage  across  the  re- 
ceiver producing  an  interfering 
effect  equal  to  that  produced  by 
the  reference  noise  voltage  at 
1000  cycles.  For  a  given  re- 
ceiver, the  voltage  for  reference 

noise  at  a  frequency  other  than  1000  cycles  may  be  obtained  from  the  1000-cycle  reference 
voltage  and  the  proper  noise  weighting  curve  for  receiver  voltages  (if  available) .  Figure  8 
shows  such  weighting  curves  for  the  three  receivers  listed  in  Table  3.  Also,  for  a  given  re- 
ceiver, the  receiver  current  for  reference  noise  at  a  frequency  other  than  1000  cycles  may  be 
obtained  from  the  relation  between  the  voltage  for  reference  noise  across  the  receiver  at  the 
given  frequency  and  the  receiver  impedance  at  the  same  frequency.  Values  of  receiver 
currents  and  corresponding  voltages  for  reference  noise  at  various  odd  harmonics  of  60 
cycles,  obtained  in  accordance  with  the  above  procedure,  are  given  in  Fig.  13. 

From  the  current  for  reference  noise  at  a  given  frequency  /  and  the  susceptiveness  factor 
at  the  same  frequency  for  a  subscriber  set,  the  receiver  noise  (in  decibels)  for  1  volt  to 
ground  on  the  set  at  that  frequency  is  given  by  the  expression 


Receivers 
(W.E. 
Go.  types) 

Current, 
micro- 
amperes 

Voltage, 
micro- 
volts 

Impedance, 
ohms 

144 
557 
HA1 

0.0795 
0.1414 
0.1210 

21.60 
19.70 
16.56 

158  +  J221  =  272/54.5° 

50  +jl30  =  139/69° 
68+jllS  =  136/60° 

Nf  (receiver  noise  in  db)  =  20 

where  I/  —  receiver  current  in  microamperes  per  volt  to  ground  at  the  subscriber  set  at 

frequency  /  =  suseeptiveness  factor  for  set  involved. 

IT  =  receiver  current  in  microamperes  (for  the  type  of  receiver  involved)  for  refer- 
ence noise  at  frequency  /  (Fig.  13  for  three  different  receivers). 

Values  of  Nf  for  the  144  receiver,  calculated  by  the  above  expression,  require  no  correc- 
tions, since  present  receiver  noise  transmission  impairments  are  related  to  noise  magnitudes 
in  this  type  of  receiver.  Values  of  Nf  so  calculated  for  the  557  and  HA1  types  of  receivers 
are  not  representative  of  the  interfering  effect  on  the  present  standard  decibel  scale  of 


NOISE  FREQUENCY  INDUCTION 


10-81 


noise  for  the  144  receiver.  For  the  final  results  of  all  three  receivers  to  be  on  approximately 
the  same  basis  as  to  noise  transmission  impairments,  it  is  necessary  to  adjust  the  JV/ 
values  of  the  557  and  HA1  receivers  by  factors  of  -f  4db  and  —  4db,  respectively,  which 
factors  are  based  on  the  results  of  judgment  tests.  By  such  adjustments,  the  readings  are 
in  dba,  a  term  that  can  be  interpreted  as  representing  the  quantity  that  results  after 
properly  "adjusting"  the  decibel  reading  of  the  noise  measuring  set.  When  the  reading  is 
thus  adjusted  all  noise  results  will  be  on  a  common  basis. 


Receiver 
>—  » 

Currents 

Voltages 

Receiver 

Currents           j          Voltages 

144 

557 

HA1 

144 

557  |HAJ 

144 

557    |  HAl 

144 

557 

HAl 

Fre- 
quency, 
cpe 

Microamperes 

Microvolts 

Fre- 
quency, 
cps 

Microamperes 

Microvolts 

180 
300 
420 

4.21 
0.954 
.586 

9.29 
2.11 
1.27 

1.41 
0.566 
.338 

530. 

144. 
106. 

483. 
131.5 
96.6 

60.8 
31.7 
23.6 

1620 
1740 
1860 

0.304 
.334 
.365 

0.505 
.549 
.586 

0.147 
.154 
.157 

112. 
128. 
146. 

101.5 
116.9 
133. 

26.2 
28.2 
29.9 

540 
660 
780 

.40 
.281 
.180 

0.82 
.542 
.34 

.240 
.181 
.149 

80.1 
60.6 
43.3 

73. 
55.3 
39.5 

20.1 
18.1 
17.1 

1980 
2100 
2220 

.373 
.376 
.377 

.604 
.598 
.593 

.158 
.156 
.154 

159. 
167. 
167. 

145. 
152. 
159. 

31.5 
32.9 
34.4 

900 
1020 
1140 

.109 
.077 
.094 

.205 
.138 
.161 

.131 
.120 
.123 

28.7 
21.2 
27.1 

26.2 
19.5 
24.7 

16.7 
16.7 
18.5 

2340 
2460 
2580 

.377 
.377 
.377 

.591 
.588 
.580 

.152 
.150 
.148 

181. 
185. 
193. 

165. 
172. 
178. 

35.8 
37.2 
38.9 

1260 
1380 
1500 

.138 
.2 
.258 

.235 
.343 
.445 

.129 
.134 
.140 

42.4 
66.2 
91.2 

38.6 
60.3 
83.6 

20.5 
22.3 
24.3 

2700 

.377 

.569 

.155 

200. 

182. 

40.8 

FIG.  13.     Receiver  Currents  and  Voltages  Corresponding  to  Reference  Noise.*    (Report  46,  J.C.P.C.) 

*  Weighted  currents  and  voltages  based  on  representative  receiver  impedances  and  definition  of 
reference  noise  as  10~12  watt  dissipated  in  each  receiver  at  1000  cycles. 

Power  Level  and  Sensitivity.  The  susceptiveness  of  telephone  circuits  to  induced  noise 
from  power-supply  circuits  or  other  outside  influences  depends  to  a  considerable  degree 
on  the  levels  of  the  voltages  and  currents  used  in  speech  transmission  and  in  the  efficiency 
of  the  telephone  terminal  apparatus  in  converting  electrical  into  sound  power.  Power 
level,  as  discussed  here,  refers  to  the  level  of  speech  currents  (with  respect  to  reference 
level  of  speech  transmission)  and  not  to  a  level  measured  in  watts. 

Speech  power,  and  consequently  electrical  power  generated  by  a  subscriber  set  telephone 
transmitter,  which  is  actuated  by  speech  power,  varies  over  a  wide  range  of  values  and 
frequencies.  This  variation,  will  occur  with  any  one  speaker  and  is  usually  different  for 
different  speakers.  Electrical  power  also  varies  in  different  toll  and  exchange  circuits, 
owing  to  the  different  types  of  lines  and  apparatus  encountered  and  line  length.  In  one 
investigation,  the  average  acoustic  power  of  speech  produced  by  16  talkers  was  of  the  order 
of  10  microwatts.  The  average  ratio  of  the  maximum  instantaneous  power  to  the  average 
power,  for  the  various  vowel  sounds  only,  was  of  the  order  of  15  to  1,  whereas  the  ratio  of 
the  maximum  to  average  power  for  a  continuous  sine  wave  is  2  to  1,  thus  showing  the 
much  wider  variation  of  speech  power  as  compared  to  generated  electric  power. 

The  power  on  commercial  telephone  circuits  is  conveniently  determined  by  a  device 
known  as  the  volume  indicator.  This  device  consists  primarily  of  a  rectifier-type  indicating 
meter  of  specific  dynamic  characteristics.  When  measuring  speech  power,  the  meter 
deflections  fluctuate  continually  in  response  to  the  variations  of  speech  power.  Because 
of  this  varying  deflection,  it  is  necessary  to  specify  a  standard  method  of  interpreting  the 
indications,  which  involves  adjusting  a  calibrated  potentiometer,  associated  with  the 
meter,  to  maintain  the  meter  needle  deflections  approximately  in  a  specified  range  on  the 
scale.  When  these  deflections  correspond  to  the  specified  reading,  with  the  volume  indi- 
cator connected  across  600  ohms,  and  the  potentiometer  is  set  at  zero,  the  power  indicated 
by  the  meter  in  the  circuit  being  measured  is  zero  YU  (volume  units). 

Since,  by  the  action  of  the  carbon  granule  transmitter,  a  relatively  large  amount  of 
electrical  power  is  controlled  by  movement  of  the  diaphragm,  the  electrical  power  delivered 
to  a  telephone  circuit  is  much  greater  (of  the  order  of  several  hundred  tunes)  than  the 
acoustic  power  delivered  to  the  transmitter.  This  amplification  is  of  value  in  maintaining 
speech  power  at  satisfactory  levels  above  induced  power  current  levels. 


10-82  TRANSMISSION  CIRCUITS 

Transmitter  developments  have  tended  to  raise  the  response  level  for  some  frequencies 
in  order  to  give  a  more  nearly  flat  frequency  characteristic  over  the  voice  range  without 
materially  increasing  the  maximum  level  output  of  the  transmitter.  Such  a  characteristic 
improves  the  articulate  qualities  of  the  speech  and  hence  effective  transmission. 

Receiver  developments  have  also  increased  receiver  efficiency  over  the  earlier  periods 
of  operation,  in  addition  to  flattening  its  response  over  the  voice  range.  However,  its 
actual  efficiency  (electrical  power  input  to  acoustic  power  output)  is  relatively  low.  The 
power  loss  in  the  receiver  at  low  frequencies,  such  as  25  and  60  cycles  and  their  lower 
harmonics,  is  much  greater  than  at  frequencies  in  the  higher  voice  range.  The  combination 
of  amplification  of  voice  currents  in  the  telephone  transmitter  with  noise  current  loss  in  the 
receiver  permits  delivering  to  the  subscriber  satisfactory  speech  power  while  keeping 
within  tolerable  limits  the  sound  levels  due  to  ordinary  amounts  of  induced  noise. 

Speech  levels  on  toll  circuits  are,  in  general,  maintained  at  about  specified  levels  by 
means  of  telephone  repeaters  (not  usually  employed  on  exchange  circuits) ,  which  amplify 
the  speech  as  well  as  any  noise  currents  and  voltages  induced  in  the  toll  circuits.  These 
repeaters  are  usually  spaced  at  suitable  intervals  on  a  toll  circuit  to  provide  the  proper 
speech  level  without  overloading  the  amplifier  and  without  causing  cross-induction  between 
adjacent  circuits,  due  to  excessive  or  inadequate  levels. 

The  trend  toward  improved  balance  of  party-line  subscriber  sets  and  some  parts  of 
central-office  circuits  permits  increasing  the  speech-to-noise  level  ratio  in  exchange  plant. 
Any  betterments  of  this  type,  within  limits,  which  increase  the  signal-to-noise  ratio  tend 
to  decrease  the  noise  effects  in  telephone  circuits. 

32.  NOISE  INDUCTION  MITIGATION 

Noise  induction  mitigation  usually  involves  careful  consideration  of  at  least  several 
of  a  large  number  of  factors,  which  may  be  broadly  classified  under  the  headings:  (a)  in- 
fluence factors;  (5)  coupling  factors;  (c)  susceptiveness  factors. 

COOPERATIVE  PLANNING  in  connection  with  the  design  and  location  of  lines  and 
systems  is  of  great  importance.  These  cooperative  plans  generally  are  directed  toward:  (a) 
coordinating  the  locations  of  lines;  (5)  incorporating  in  the  design  of  both  systems  those 
features  which  will  limit  the  influence  and  susceptiveness. 

By  cooperative  planning,  not  only  can  the  number  of  exposures  be  limited  but  also  the 
general  designs  of  the  systems  can  be  made  such  that  treatment  of  individual  exposures 
is  materially  simplified.  Furthermore,  in  connection  with  new  construction  or  changes 
in  either  system,  coordination  can  be  considered  before  expenditures  or  other  commitments 
are  made. 

CONTROL  OF  INFLUENCE  FACTORS.  Residual  currents  and  voltages  of  the 
triple-harmonic  series  can  be  controlled  by  one  or  more  of  the  following  means : 

(a)  Opening  the  neutral-to-ground  connection  of  the  machine  or  transformer  bant 
where  the  triples  originate.  This  can  be  done  only  where  other  system  grounding  arrange- 
ments, adequate  from  the  standpoint  of  power  system  stability  and  relaying,  are  available. 

(6)  Opening  the  neutral-to-ground  connection  of  transformer  banks  through  which 
triples  from  another  source  complete  their  path.  System  stability  and  relaying  must  also 
be  considered  in  this  connection. 

(c)  Providing  a  path  for  triples  (such  as  with  a  wye-delta  bank),  which  tends  to  shunt 
them  out  of  the  exposure. 

(d)  The  use  of  wave  traps  (anti-resonant  circuits  tuned  to  the  important  harmonics), 
reactances,  or  other  devices  in  the  neutral-to-ground  connections  at  locations  where  triples 
originate. 

(e)  The  use  of  transformer  connections  (such  as  wye-delta  or  delta-delta)  through 
which  triples  will  not  pass. 

(/)  The  use  of  rotating  machines,  which  have  a  low  influence  factor. 

Note:  The  last  two  measures  are  usually  of  the  greatest  importance  in  connection  with  cooperative 
advance  planning,  since  to  change  existing  equipment  to  these  types  may  be  unduly  expensive. 

Residual  currents  and  voltages  of  the  non-triple  harmonic  series  can  be  controlled  by: 

(a)  Reduction  of  the  unbalance  which  gives  rise  to  them,  as  by  changing  single-phase 
taps  to  3-phase,  balancing  single-phase  taps  among  the  phases  of  the  3-phase  line,  or 
balancing  loads  among  the  3  phases,  where  neutrals  are  multigrounded. 

(6)  Absorption  by  wye-delta  banks  or  other  means. 

In  addition  to  the  design  of  power  apparatus  to  limit,  harmonics  as  far  as  practicable, 
and  the.  avoidance  of  excessive  magnetic  densities,  frequency  selective  devices,  to  filter 
o*xt  harmonics,  have  been  used  in  some  situations,  for  example: 


NOISE  INDUCTION  MITIGATION  10-83 

(a)  On  the  d-c  sides  of  trolley  rectifiers, 
(fc)  On  the  a-c  sides  of  rectifiers. 

(c)  Across  the  terminals  of  rotating  machines  to  reduce  important  balanced  harmonic 
currents  and  voltages. 

(d)  In  the  neutral-to-ground  connections  of  generators,  synchronous  converters,  and 
other  power  devices,  to  reduce  triple-harmonic  voltages 

and  currents,  an  example  of  which  is  shown  in  Fig.  14. 

CONTROL  OF  COUPLING  FACTORS.  One  highly 
satisfactory  method  for  the  control  of  inductive  cou- 
pling, when  it  can  be  employed,  is  the  complete  physical 
separation  of  the  power  and  telephone  lines.  However, 
in  built-up  communities  both  types  of  service  are  re- 
quired by  the  public,  making  it  necessary  to  utilize  the 
same  routes  for  distribution.  For  intercity  toll  lines,  ~±r 

which  are  usually  important  backbone  routes  for  long- 
haul  communication  traffic,  frequently  reasonable  sepa-     SUp^ency  wMch  trap  b  tUBed  to 
ration  from  power  transmission  systems  can  be  ob-  103 

tained,  particularly  with  the  proper  cooperative  advance  ~~      ~ 

planning. 

Where  power  line  parallels  with  open-wire  communi- 
cation lines  (toll  or  exchange)  are  created,  transpositions 
are  usually  effective  in  controlling  the  resulting  indue-  I?G-  14,-  w.ave  TraP  *&  Grounded 
tive  couplings,  and  these  may  be  required  on  a  coordi-  eu  (Repo^S?j!c!p.C Jnerator 
nated  basis. 

Whether  the  coordination  of  telephone  transpositions  with  the  discontinuities  in  the 
exposure,  or  the  use  of  power-circuit  transpositions,  or  both,  is  desirable  in  a  specific  case 
will  depend  on  the  relative  importance  of  direct  metallic-circuit  induction  and  longitudinal- 
circuit  induction  acting  on  telephone-circuit  unbalances,  and  on  the  importance  of  the 
induction  from  the  balanced  and  residual  components  of  the  power-circuit  voltages  and 
currents. 

Telephone  transpositions  must  also  be  effective  in  controlling  cross-induction  (cross- 
talk) between  telephone  circuits.  Standard  transposition  arrangements  have  been  devised 
to  meet  this  requirement  for  different  classes  of  open-wire  facilities.  Two  of  such  arrange- 
ments, which  are  available,  are  shown  for  four  arms  of  wire  in  Figs.  15  and  16. 

Sometimes,  unavoidable  irregularities  occur  in  the  spacing  of  poles,  in  distances  between 
power  and  telephone  circuits,  in  the  presence  of  shielding  objects  such  as  other  communi- 
cation lines  and  trolley  systems,  and  in  heights  of  the  circuits,  which  it  is  not  practicable 
to  take  into  account  in  the  transposition  design.  Where  these  irregularities  are  large,  the 
effectiveness  of  the  transposition  arrangements  is  correspondingly  impaired. 

Possible  benefits  are  illustrated  by  noise  measurements  in  the  rural  joint-use  exposures, 
established  in  Alabama,  Minnesota,  and  South  Dakota  (see  article  29) .  The  ratio  of  re- 
ceiver noise  to  noise-to-ground  is  a  good  indicator  of  the  effectiveness  of  noise-reduction 
measures,  and  such  measurements,  made  under  normal  conditions,  gave  ratios  of  the  order 
of  —  36  db  (about  1  to  60  in  voltage) ,  where  the  IT  (effective  power  harmonic  currents  in 
the  line  times  the  voltage  interference  factor)  is  fairly  high.  A  ratio  of  this  order  is  ade- 
quate with  respect  to  noise,  except  in  extreme  exposures,  and  should  result  in  noise  on 
rural  joint-use  circuits  comparable  to  that  on  urban  party-line  circuits  in  cable. 

The  favorable  noise  results  in  these  instances  are  due  partly  to  the  effectiveness  of  the 
transposition  scheme  used,  which  employs  frequent,  point-type  transpositions  on  tandem 
brackets  and  an  average  wire  spacing  of  only  about  7  in.  Also  in  many  of  the  exposures 
the  power  circuit  is  of  the  single-phase  common-neucral  type  with  vertical  configuration, 
for  which,  in  joint  use,  the  direct  metallic  induction  into  a  horizontal  telephone  circuit  is 
inherently  low.  Of  course,  low  values  of  receiver  noise  are  not  obtainable  unless  the 
telephone  circuits  are  well  balanced. 

Shielding.  When  either  or  both  of  the  telephone  and  power  facilities  are  enclosed  in 
metallic  cable  sheath,  having  a  relatively  low  resistance  to  ground,  the  sheath  acts  as  an 
effective  shield  against  both  electric  and  magnetic  induction.  For  magnetic  induction,  the 
induced  longitudinal  currents  fiow  along  the  sheath,  which  has  a  finite  resistance  per  unit 
of  length,  different  from  the  enclosed  conductors,  and  complete  shielding  from  such  cur- 
rents is  thus  not  obtainable. 

Present  trends  are  toward  bonding  local  distribution  aerial  cable  to  multi-grounded 
power  neutrals,  where  these  neutrals  are  well  bonded  to  extensive  public  water  systems, 
since,  in  the  usual  case,  the  cable  sheath  becomes  quite  closely  associated  with  the  common 
power  neutral  through  the  telephone  drops,  station  protector  grounds,  power  secondary 
services,  and  telephone  company  practices  of  placing  protector  blocks  between  working 


10-84 


TRANSMISSION  CIRCUITS 


lines  and  the  cable  sheath.  Frequent  bonds  between  the  common  neutral  and  cable 
sheath  (about  every  x/4  mile  or  less)  are  not  only  advantageous  from  a  protection  stand- 
point but  also  useful  for  noise  mitigation. 


Such  bonding  usually  results  in  an  average  reduction  of  subscriber  receiver  noise  of  the 
order  of  2  to  1  for  station  sets  equipped  with  single  condensers  and  low-impedance  ringers  to 
ground.  The  average  reduction  in  noise  to  ground  is  about  3  to  1 .  In  non-public  water-pipe 
areas  the  noise  reductions  obtained  by  bonding  to  the  neutral  depend  upon  the  resistance 
to  ground  of  the  neutral  conductor:  the  lower  the  resistance,  the  greater  the  reduction. 

For  power  systems  of  the  2.3/4.0-kv  type,  increased  potentials  on  the  telephone  sheath 
may  generally  result  from  bonding  to  the  power  neutral,  but  tests  indicate  no  cable  iusiila- 


NOISE  INDUCTION  MITIGATION 


10-85 


r   1.2 

1st  |  3-4 
7-8 
9-10 


Arm  } 


2nd 
Arm 


3rd  ( 
Arm" 


4th  ^ 
Arm 


'  .11-12 
13-14 

17-18 
,  19-20 

'  21-22 
23-24 

27-28 
.  29-30 

'  31-32 
33-34 

37=38 
.  39-40 


5-6 

15-16 

25-26 
35-36 

45-46 
55-56 

65-66 

75-76 

8        16       24      32      40      48      56      64      72      SO       88      96     104    li2    120 
NOTE: — The  figure  at  each  phantom  transposition  indicates  the  type  (see  Fig.  11). 

FIG.  16.    Typical  Transposition  Scheme  (Improved  Type)  for  Phantomed  Circuits  Suitable  for  Use 
in  Inductive  Exposures  (Courtesy  Bell  System) 

tion  failures  due  to  this  potential  increase.    In  trolley  areas,  additional  direct  current  will 

usually  be  transferred  to  the  underground 

telephone   cable   sheaths   because   of  power 

neutral  bonding,  but  the  increase  in  sheath 

current  will  probably  not  require  additional 

corrective  measures  for  electrolysis. 

The  effectiveness  of  shielding  is  expressed 
in  terms  of  a  shield  factor,  which  is  the  ratio 
of  the  noise  in  the  shielded  to  the  noise  in  the 
non-shielded  condition.  Figure  17  shows  ob- 
served shield  factors  in  four  public  water 
system  areas  obtained  with  telephone  cable 
sheaths  bonded  to  a  multigrounded  power 
circuit  neutral  at  a  number  of  places. 

CONTROL  OF  SUSCEPTIVENESS.  In 
toll  circuits,  which  are  designed  to  be  sym- 
metrical with  respect  to  earth,  the  reduction 
of  unbalances  is  usually  a  matter  of  correcting 
conditions  which  are  the  result  of  deteriora- 
tion or  maintenance,  although  situations 
occasionally  arise  where  the  design  of  appa- 
ratus is  involved.  The  former  includes: 

(a)  High-resistance  joints.  The  remedy  is 
to  make  a  new  joint. 


^ 

**= 

== 

S52- 

*•—  « 

\ 

s 

\ 

v 

NOISE  TO 
vGROUNO 
t 

^ 

V 

I 

t 

RECEIVER  NCHSE,! 
PARTY   UNE  SET  > 
1-SA  RINSE*   TO 

N 

I 

( 

\ 

V 

\ 

-v^ 

\ 

^ 

\ 

SHIELD  FACTOR 


OL3        CL2        OJ 


FIG.  17.    Shield  Factors  —  Telephone  Cable  Sheath 


^«^  «  ^cw  JW1I1U.  ...       Bonded  to  Power  Circuit  Neutral  (Courtesy  Bell 

(o)  Leakage  caused  in  open-wire  circuits  System) 


10-86 


TRANSMISSION  CIRCUITS 


by  trees  or  broken  or  missing  insulators.    In  cable  circuits,  leakage  is  usually  caused  by 
moisture  entering  at  a  sheath  break. 

(c)  Capacitance  unbalances  in  open-  wire  circuits.    The  remedy  involves  a  careful  check 
of  the  transpositions  either  by  inspection  or  by  suitable  electrical  testing  means. 

(d)  Incorrect  connections  or  unbalanced  arrangements  of  apparatus  at  terminals.    For 
example,  composite  sets  should  not  be  placed  on  one  side  circuit  of  a  phantom  group  with- 
out similar  sets  being  placed  on  the  other. 

(e)  Incorrect  connections  in  entrance  cables,  such  as  split  pairs  or  quads. 

In  regard  to  design,  the  causes  of  unbalances  are  usually  in  terminal  apparatus,  where 
the  elements  in  the  two  wires  of  a  circuit  (or  the  two  sides  of  a  phantom  group)  are  not 
sufficiently  alike  in  impedance  at  noise  frequencies.  Modern  apparatus  is  usually  designed 
so  that  the  series  impedances  and  admittances  to  ground  are  very  closely  similar  for  the 
wires  in  a  pair  or  quad.  In  some  of  the  older  designs,  however,  the  degree  of  balance 
may  not  be  sufficiently  high.  Sometimes,  improvements  can  be  secured  by  selecting  among 
existing  equipment  the  units  having  similar  characteristics  and  grouping  them  together 
on  pairs  or  quads.  In  some  instances,  there  may  be  unbalances  in  entrance  cables  or 
office  cabling,  for  example  where  phantomed  circuits  are  routed  through  non-quadded 
cables. 

In  the  exchange  plant,  unbalances  due  to  connections  of  ringers  to  ground  can  be  reduced 
by  the  use  of  high-impedance  ringers  or  by  other  subset  apparatus  with  improved  balance. 
Central-office-cireuit  unbalances  may  need  to  be  improved  by  modifications  in  or  replace- 
ments of  existing  apparatus  of  the  older  types  and  of  unsymmetrical  design.  The  more 
recently  designed  central-office  equipment  is  better  balanced,  and  further  improvements 
in  this  respect  may  be  expected. 

Sometimes  improvement  can  be  secured  by  inserting  a  balancing  impedance  in  the  other 
side  of  the  circuit.  Through  cooperative  advance  planning,  apparatus  having  improved 
balance  can  be  introduced  in  an  orderly  manner. 

Isolation  of  equipment  unbalances  can  sometimes  be  secured  by  inserting  between  the 
apparatus  and  the  line  a  well-balanced  repeating  coil  without  ground  connections  or  with 
ground  connections  so  arranged  that  longitudinal  voltages  and  currents  are  not  trans- 
mitted. It  is  necessary  to  arrange  the  circuit  so  that  signaling  and  supervision  will  not 
be  interfered  with.  A  less  effective  but  sometimes  adequate  method  of  isolation  consists 
of  inserting  between  the  apparatus  and  the  line  a  well-balanced  coil  so  connected  as  to  be 
non-inductive  to  the  metallic  circuit  but  to  present  a  high  longitudinal  impedance.  A 
well-balanced  repeating  coil,  with  the  windings  suitably  connected,  will  frequently  serve 
this  purpose.  This  method  has  the  advantage  that  it  can  be  readily  arranged  so  as  not  to 
interfere  with  d-c  signaling  and  supervision.  In  both  the  toll  and  exchange  plants,  it 
is  frequently  necessary  to  guard  against  interconnection  of  balanced  and  unbalanced 
circuits  through  cord  circuits  not  containing  repeating  coils,  since  such  a  connection  would 
be  unbalanced.  This  can  be  done  by  avoiding  the  use  of  such  cord  circuits  for  these  connec- 
tions or  by  isolating  tbe  unbalanced  lines  by  repeating  coils. 

Figure  18  shows  a  schematic  of  a  typical  local  step-by-step  connector  circuit  in  the 

talking  condition,  with  a  long 
UOK6  UKE  cmcurr  LOCAL  coKNECTQ*  Hne  ^^  inserted  adjacent 

R  to  the  line  in  order  to  prevent 
longitudinal  current  flow  from 
the  calling  line  causing  noise, 
due  to  possible  unbalances  in 
relays  A  and  D  and  in  the  2- 
yuf  series  condensers.  This  is 
one  means  of  preventing  lon- 
circuit  noise  reach- 
called  subscriber  from 
a  connected  line  which  is 
affected  by  power  induction. 

Transmission  Levels.  The  effect  of  noise  is  lowered  as  the  power  level  of  the  voice 
currents  of  telephone  circuits  is  raised.  In  the  toll  plant,  this  fact  has  had  a  marked 
bearing  on  the  sizes  of  wire  used  and  the  location  of  repeater  stations.  One  of  the  limita- 
tions on  the  degree  to  which  levels  can  be  raised  on  toll  circuits  by  repeaters  is  the  difficulty 
of  avoiding  cross-talk  between  circuits  on  which  there  are  large  level  differences.  Subject 
to  this  limitation,  however,  advantage  may  sometimes  be  taken  in  specific  situations  of 
allocating  repeater  gains  in  such  a  way  as  to  use  the  highest  practicable  level  through 
inductive  exposures. 

In  the  exchange  plant,  telephone  repeaters  are  normally  not  employed,  so  that  the  con- 
trol of  levels  in  connection  with  specific  noise  situations  is  a  less  practicable  procedure. 


ZMF 


FIG.  18.     Schematic  of  Typical  Local  Step-by-atep  Connector  Cir- 

euit  in  Talking  Condition  Associated  with   a  Long  Line  Circuit 

(Courtesy  Bell  System) 


NOISE  INDUCTION  MITIGATION 


10-87 


However,  the  desirability  of  utilizing  the  highest  practicable. levels  has  had  an  important 
bearing  in  the  development  of  instruments,  cables,  and  other  facilities. 

Other  Devices.  In  special  cases,  neutralizing  transformers,  resonant  shunts,  or  resonant 
drainage  to  ground,  applied  to  the  telephone  circuits,  offer  possibilities  as  coordinative 
measures  for  the  reduction  of  noise  induction. 

The  neutralizing  transformer  is  employed,  primarily,  in  local  communication  circuits 
serving  power  stations  to  limit  voltages  to  ground  at  such  stations  when  power-line  faults 


ACCIDENTAL 
GROUND 


-NOTE  —  X 

IN  A  PRACTICAL  SITUATION     X. 
THE  EQUIPOTENTIAL  LINES 
WOULD  NOT  BE  AS  REGULAR 
AS  INDICATED 


*  FIGURES  USED  AS 
ILLUSTRATION  ONLY 


FIG.  19. 


Diagram  Illustrating  Rise  in  Station  Ground  Potential  Due  to  Power  Line  Fault  (Report  44, 

J.C.P.C.) 

occur.  Without  the  transformer,  the  rise  in  potential  of  the  power  station  ground,  and 
consequently  the  telephone  set  protector  ground,  would  frequently  be  enough  to  break 
down  the  protector  blocks  and  disable  telephone  service,  as  shown  in  Fig.  19.  With  the 
transformer,  differences  of  potential  between  the  communication  circuit  and  nearby 
grounded  structures  are  materially  reduced,  as  indicated  in  Fig.  20,  thus  minimizing 
hazard  to  personnel,  cable  troubles,  and  service  interruptions. 


PROTECTOR                 r      V->    "*                  PROTECTOR 

BLOCKS    ^ 

-^TOT^ 

BLOCKS 

f                                         6X^ 

1                                     f 

I 

TO               \     (3) 
SUBSCRIBER          KVo—w 
SHIPMENT        /         R 

C±D 

^ 
^b 

|     ^~[ 

•j 

R\ 

TO 
CENTRAL 
OFFICE 
EQUIPMENT 

/ 

^ 

1          f<~\    r^,, 

v? 

-nwr*- 

T 

\ 
\ 

Tf 

*        o  VJD. 

Vp 

rCJL/4  •"  v.yy    '"Yn^"vvv 

"S  PQWER- 
— L_  STATION 
—  GROUND 


""f    CENTR 
— I—    OFFIC 


CENTRAL- 
OFFICE 
~     GROUND 


^  GROUND    ~ 

(T)    CONNECTION   MADE  VIA  CABLE  SHEATH   WHEN   AVAILABLE. 

(?)    USE    REMOTE  GROUND  WHEN    NO   CABLE  SHEATH   IS   AVAILABLE. 

(3)    REMANENT  VOLTAGE    (VR)  =  V,-V2.  "K*  WILL  HAVE  A  VALUE 

BETWEEN    1/2   AND    t    DEPENDING  UPON   CIRCUIT  CONDITIONS. 

FIG.  20.     Diagram  Illustrating  Action  of  Neutralizing  Transformer  in  Neutralizing  Voltages  on  a 
Circuit  Subject  to  Ground  Potential  Rise  (Report  44,  J.C.P.C.) 

Ground  Return  Circuits.  Ground  return  telephone  circuits,  being  inherently  unbalanced 
to  ground,  require  methods  of  coordination  with  power  circuits  different  from  those  em- 
ployed for  metallic  telephone  circuits,  such  as  previously  discussed  in  this  section.  Ground 
return  power  circuits  are  not  considered  standard  practice  but  are  frequently  operated 
as  single-phase  grounded  neutral  circuits  in  rural  power  systems.  Coordinative  measures 
are  available  for  this  type  of  power  circuit  with  metallic  telephone  circuits,  such  as  control 
of  power-circuit  influence  factors  and  transpositions,  drainage  coils,  or  high  impedances 
(to  predominating  harmonics)  inserted  in  the  telephone  circuit. 


10-88 


TRANSMISSION   CIRCUITS 


33.  LOW-FREQUENCY  INDUCTION 

Low-frequency  induction,  under  normal  balanced  power  system  operation,  rarely  creates 
a  noise  problem  in  communication  systems.  However,  under  some  normal  operating 
conditions,  the  induced  ground- wire  (for  lightning  protection)  current  flow  at  the  power- 
circuit  fundamental  frequency  may  be  objectionable,  particularly  in  telegraph  operation. 
When  an  abnormal  condition,  such  as  a  grounded  phase  wire,  occurs  on  a  power  circuit, 
relatively  large  currents  at  the  fundamental  frequency  flow  from  the  power  circuit  to 
ground  for  grounded  power  systems.  This  may  result  in  excessive  low-frequency  induction 
in  any  paralleling  communication  circuits.  This  condition  also  obtains  in  an  isolated  power 
system  when  two  or  more  phases  are  faulted. 

The  magnitude  of  the  voltage  induced  in  a  telephone  circuit  at  the  time  of  a  power- 
circuit  fault  depends  chiefly  on  the  magnitude  of  residual  currents  and  on  the  exposure 
conditions. 

Residual  Currents.  If  1  phase  of  a  3-phase  power  line  develops  a  fault  to  ground  the 
currents  in  the  3  phases  become  unequal,  and  their  vector  sum,  which  is  the  residual 
current,  is  no  longer  zero.  In  most  low-frequency  induction  problems,  residual  current 
is  far  more  important  than  residual  voltage. 

The  relatively  large  inductive  influence  of  residual  current  is  due  to  the  fact  that  it 
exists  in  a  circuit  consisting  of  the  line  conductors  in  parallel  as  one  side  and  the  earth 
as  the  other  side.  Since  much  of  the  return  current  is  deep  in  the  earth,  its  neutralizing 
action  is  small. 

The  chief  factors  that  determine  the  magnitude  of  the  residual  currents  are  (1)  the  power- 
circuit  voltage,  (2)  the  line  and  apparatus  impedances,  (3)  the  fault  and  earth  impedances, 
(4)  the  impedances  of  the  neutral  ground  connections,  (5)  the  type  of  ground  wire,  if  used, 
and  (6)  the  circuit  configuration  including  ground  wires. 

In  analyzing  the  impedances  controlling  fault  current,  two  general  types  of  power 
systems  must  be  considered:  the  grounded  neutral,  and  isolated  neutral  systems.  These 

are   illustrated   in   Fig.    21.     In  the 

Transmission  Line Transformer_     grounded  neutral  system,  the  neutrals 

of  one  or  more  transformer  banks  are 
grounded  directly  or  through  imped- 
ance so  that,  in  the  event  of  a  fault,  a 
path  for  current  is  established  from 
the  fault  through  the  earth  and  return- 
ing through  the  neutral-to-ground  con- 
nections. In  the  isolated  neutral  sys- 
tem, there  are  normally  no  grounds  on 
the  system  so  that,  in  the  event  of  a 
fault,  the  path  for  fault  currents  is 
through  the  capacitances  of  the  un- 
faulted  phases  to  earth.  Hence,  for  a 
single  fault,  the  fault  current  is  limited 
to  the  charging  and  leakage  current. 
If,  in  an  isolated  neutral  system,  a 
second  fault  to  ground  occurs  on 
another  phase  while  the  first  persists,  a 
residual  current  will  flow  between  the  faults.  Simultaneous  faults  on  two  phases  at 
different  points  may  occur  on  any  type  of  system  but  are  more  likely  to  occur  on  an 
isolated  neutral  system  than  on  one  in  which  the  neutral  is  solidly  grounded.  This  is 
because,  for  the  isolated  system,  full  phase-to-phase  or  possibly  higher  voltage  is  impressed 
between  the  unfaulted  phases  and  ground,  thus  increasing  the  voltage  stress  on  the  insu- 
lation of  the  entire  system  during  the  time  of  fault. 

A  system  grounded  through  a  neutral  impedance  has  some  of  the  characteristics  of  an 
isolated  neutral  system.  Generally,  the  addition  of  neutral  impedance  tends  to  reduce 
the  fault  current,  this  effect  being  proportionately  larger  for  faults  near  the  neutral  ground- 
ing points.  This  reduces  the  voltage  induced  on  nearby  telephone  lines  and  may  have 
advantages  to  the  power  system.  On  the  other  hand,  increasing  the  neutral  impedance 
may  introduce  problems  in  power  system  relaying.  It  may  also  increase  overvoltages  on 
the  power  system. 

The  duration  of  residual  current  is  also  important,  since  the  length  of  time  that  the 
induced  voltage  persists  has  important  reactions  on  its  effects.  For  example,  with  the 
carbon  block  protectors,  the  chance  of  their  becoming  permanently  grounded,  with  con- 
se<nient  interruption  of  service,  depends  not  only  upon  the  amount  of  current  through 


(a)  Isolated  System  with  Delta  Connections 


Supply 
Transformer 


Transmission  Line 


Load 
Transformer 


(6)  Grounded  System  with  Delta-wye  Connections 

FIG.  21.    Power  System  Arrangements  with  Isolated  and 
Grounded  Connections 


LOW-FREQUENCY  INDUCTION 


10-89 


the  protector  but  also  upon  its  duration.  Likewise,  other  effects  which  are  described  later 
are  materially  affected  by  tl^e  duration  of  the  induced  voltage.  Since,  except  for  self- 
clearing  faults,  the  duration  of  fault  current  is  determined  by  the  time  of  operation 
of  the  power-current  interrupting  devices,  their  reliability  and  speed  of  operation  are 
important. 

COUPLING  FACTORS.  Coupling  is  proportional  to  the  length  of  the  exposure  for 
uniform  separation.  It  varies  with  separation  in  a  manner  which  is  affected,  among  other 
things,  by  the  structure  of  the  earth.  This  effect  can  be  summarized  as  follows: 

Under  the  conditions  of  low-frequency  induction,  the  telephone  wires  comprise  one  side 
of  a  loop,  the  other  side  of  which  is  the  earth.  Likewise,  the  power  wires  comprise  one  side 
of  a  loop,  the  other  side  being  the  earth.  The  magnetic  coupling  between  two  parallel 
loops  at  a  given  separation  increases  as  the  size  of  the  loops  increases.  The  sizes  of  the 
loops  are  determined  by  the  distribution  of  the  return  current  in  the  earth.  Generally 
speaking,  the  greater  the  resistivity  of  the  earth,  the  more  the  current  will  spread  and  the 
greater  will  be  the  coupling  to  an  adjacent  circuit. 

The  effect  of  earth  resistivity  on  coupling  is  much  greater  for  wide  separations  than 
where  the  lines  are  close  together.  Consequently,  with  high-resistivity  earth,  not  only  is 
the  coupling  higher  at  all  separations  than  with  low-resistivity  earth,  but  (except  for  very 
wide  separations)  the  percentage  reduction  secured  by  increasing  the  separation  a  given 
amount  is  smaller. 

Figure  22  shows,  for  several  earth  resistivities,  the  variation  of  mutual  resistance, 
reactance,  and  impedance  with  separation,  based  on  calculations  using  Carson's  formula 
and  a  frequency  of  &0  cycles,  without  shielding. 


gO.OOO* 


s 


NOTE:  SHIELDING  EFFECTS  NOT  A  FACTOR 

IN  COMPUTING  ABOVE  CURVES. 

MUTUAL    IMPEDANCE 

-——>—--—  MUTUAL    RESISTANCE 

MUTUAL    REACTANCE 

HEIGHT  OF   POWER  LfNE,  hi  =  50  FT 
HEIGHT   OF  TELEPHONE;  LINE,  H2  =25  FT 
EARTH   RESISTIVITY    p,  IN   METER^OHMS 


10 


40     60 


KX>         20O         4OO  6OO     IgOOO      2XKX5     4«OOQ          10<000    20.000 
HORIZONTAL  SEPARATION   IN   FEET 

FIG-  22.     Variation  of  Mutual  Resistance,  Reactance,  and  Impedance  with  Separation  (at  60  cycles) 

(Report  14,  J.C.P.C.) 

SHORT-CDR.CTJIT  CURRENTS,  during  power-line  faults,  may  be  calculated  by  several 
approximate  short-cut  methods,  one  of  which  may  briefly  be  described  as  follows: 

(a)  Prepare  a  diagram  of  the  power-system  network,  showing  lengths,  location,  and 
kva  capacities  of  large  generating  sources  and  transformer  banks;  location,  kva  capacities, 
and  connections  of  grounded  neutral  transformer  banks;  and  magnitudes  of  neutral  im- 
pedances, if  any. 

(b)  Show  the  location  and  separation  of  the  telephone-circuit  exposure  or  exposures  to 
the  power-system  network. 

(c)  Show  the  ]ine-to-line  operating  power  voltage. 


10-90 


TRANSMISSION   CIECUITS 


too 
so 


§40 


GENER-J  STEAM  TURBO  l_    _ 
ATORS  \HYDRO  DRIVEN  35%  A 
SYNCHRONOUS 
CONDENSERS  35% 


OjOI      CLO2      O^4  O.J 
(UNE-KV>2 


CONNeCTED  TRANSFORMER  KVA 


0,4  0.6     tO         2.  468  10         20         406000 

CONNECTED  TRANSFORMS] 
GROUNDED  ' 


3  TRANSFORMER   KVA 
TRANSFORMER  KVA 


too 

80 


RANGE  TYPE  OF 

IN  KV      GROUND  WIRE 
t       tt-33  1 

2  44-66    I  NONE  OR  LOW 

3  75- HO    f  CONDUCTIVITY  . 

4  I32-220J 


*    ^A-Wk 
5    44-220 


HKSHCON- 
- 


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2        4    e  a  10      20 

^ACTUALLY.  IMPEDANCE  BUT  USED 
AS  REACTANCE 


\ 


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J\ 


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W) 


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40   6O     1OO       20O      4OO      £00 


CALLOW  FOR  NEUTRAL  AND   FAULT  IMPEDANCE 
AS  INDICATED  JN  TEXT 


FIG.  23.     Curves  for  a  Quick  Approximation  of  Fault  Current  Resulting  from  Single-phase  Faults 
to  Ground  on  Three-phase  Systems  (Courtesy  Bell  System) 

(<3)  Using  the  various  curves  a,  b,  c,  and  d  in  Fig.  23,  determine  for  a  given  situation : 

i     -c-    f  (line  kv)2  _ 

1.  Xj.  from  ; — ; using  Fig.  23a. 

(connected  transformer  kva  at  source  supplying  fault) 

0    _     _  connected  transformer  kva  _. 

2.  F  -  XA.  from 7—7— —  ,  usuig  Fig.  236. 

grounded  transformer  kva 

Note:  Connected  transformer  kva  and  grounded  transformer  kva  refer  respectively  to  (1)  total  con- 
nected capacity,  irrespective  of  transformer  connections,  and  (2)  transformer  banks,  which  will 
pass  zero  sequence  current. 

3-  XL  —  apparent  reactance  (actually  impedance)  of  equivalent  power  line  mileage 
from  source  to  fault.  (Use  actual  mileage,  if  only  one  power  circuit  is  involved),  using 
Fig.  23c. 

4.  X  =  F  -  XA  +  XL. 


5.  Z  —  V.R2  -f  X2,  where  R  =  neutral  and  fault  resistances,  if  any. 

6.  Short-circuit  (fault)  current  //,  using  Fig.  23<2. 

* 

Notes:  If  connected  generator  kva  is  more  or  less  than  the  connected  transformer  kva  by  a  factor  of 
2  or  more,  use  connected  generator  kva  supplying  fault,  for  Figs.  23a  and  b. 
If  a  reactance  JTjv  is  used  in  the  grounding  neutral,  replace  X  with  X  +  X&. 
Figures  23a,  c,  and  d  are  based  on  the  following  assumptions: 

1.  A  single  60-cycle  generating  station. 

2.  Radial  transmission  line. 

3.  Total  connected  transformer  kva  at  station  equals  connected  generator  kva. 

4.  Line  side  neutrals  of  all  transformer  banks  are  grounded  and  will  pass  zero  sequence  current. 

5.  All  transformer  and  generator  capacities  are  so  bussed  as  to  be  effective  in  supplying  faulted  line. 

6.  All  transformers  have  S  per  cent  reactance,  based  on  kva  rating  of  bank. 

7.  Representative  average  conductor  spacings  and  sizes. 

8.  Earth  resistivity  of  100  meter-ohms  (in  the  case  of  Fig.  23c). 

The  above  ^procedure  is  not  to  be  considered  an  exact  method  of  short-circuit  current 
calculation,  since  it  usually  provides  only  a  rough  determination,  the  method  of  symmet- 
rical components  being  the  proper  one  for  more  accurate  results  (see  Electrical  Engineers 
Handbook,  Vol.  1,  Electric  Power,  H.  Fender  and  Wm.  A.  Del  Mar,  John  Wiley  &  Sons). 


LOW-FREQUENCY  INDUCTION 


10-91 


Having  determined  the  mutual  impedance  ZM-,  and  short-circuit  current  I/,  for  any 
given  situation  and  power  line  fault,  the  longitudinal  induced  voltage  J?/,  on  the  adjacent 
communication  aerial  open  wires  or  cable,  is  given  by  the  expression 

Ef  =  If  •  ZM-  TJ 
(Ztf  from  Fig.  22  or  similar  curves;  77  =  shield  factor.) 

SHIELDING.  Another  important  factor  in  determining  the  net  coupling  is  the  effect  of 
grounded  wires  or  other  linear  grounded  metallic  structures  along  the  exposure.  Voltages 
are  induced  in  such  grounded  metallic  structures  in  the  same  way  as  they  are  induced  in 
telephone  wires,  and  these  voltages  establish  currents.  The  induced  currents  flowing  in 


MESSENGER    SIZE 
25  M  1/2*  DIA.  WM   7/16*  DlA.      JOM  3/8*  DIA, 


-f-=END  GROUND  RESISTANCES  PER 
*      UNIT  OF  CABLE    LENGTH ;  OHMS/KF 


0> 


& 


X 


X 


X 


X 


X 


TO  GROUND 

TO  SHEATH 


O.O6 


003      Of  O.15          O2  O.3  0.4       O.5     O.8  O.S        LO  J.5 

D-C  RESISTANCE  OF  SHEATH  IN  OHMS  PER  KJLOFOOT 
*SHICLD  BACTORS  ARE  FOR  VOLTAGES  APPEARING  AT  ENDS  OF 

AN  AEtfUfc  CABLE    WHICH  IS  GROUNDED  ONLY  AT  THE  ENDS, 

FIG.  24.     Shield  Factors  for  Asrial  Lead  Cable  at  60  Cycles  (Report  48,  J.C.P.C.) 

these  structures  set  up  magnetic  fields,  which  generally  oppose  those  from  the  power  wires 
and  reduce  the  induction  in  the  telephone  circuit.  The  effect  of  such  currents  in  grounded 
structures  is  known  as  shielding. 

The  magnitude  and  phase  relation  of  the  current  in  a  grounded  conductor,  and  thus  the 
shielding  provided  by  it,  depend  on  the  impedance  of  the  conductor  with  earth  return. 
Hence,  the  shielding  is  increased,  when  the  resistance  of  the  conductor  and  its  ground 
connections  is  reduced.  Metallic  cable  sheaths  comprise  an  important  type  of  shielding 
conductor,  as  shown  in  Fig.  24  for  aerial  cables  with  plain  lead  sheath  at  60  cycles;  R  is 
the  sum  of  ground  connection  resistances  at  both  ends  of  the  cable  sheath  (in  ohms)  and 
S  is  the  length  of  cable  circuit  between  terminals  or  grounds  (in  kilofeet).  With  increasing 
end-ground  resistance,  the  shield  factors  for  voltages  measured  to  ground  increase,  while 
the  shield  factors  for  the  voltages  to  sheath  (between  sheath  and  conductors)  decrease. 

The  earth  resistivity  was  assumed  to  be  100  meter-ohms.  Figure  25  shows  shielding 
effects  at  60  cycles  from  grounded  telephone  open  wires. 

In  analyzing  the  distribution  of  induced  voltage  between  a  telephone  circuit  and  ground, 
assume  first  that  no  protectors  are  operated.  Under  this  condition,  the  voltages  to  ground 


10-92 


TRANSMISSION   CIRCUITS 


FREQUENCY*  eO  CYCLES 
EARTH  RESISTIVITY  = 
tOO  METER -OHMS 


R=SUM  OF  RESISTANCES  OF  END 
GROUND  CONNECTIONS  IK   OHMS 

U=  DISTANCE  BETWEEN  END  GROUND 
CONNECTIONS  IN  KJLOFEET 


on  the  telephone  wires  at  various  points  are  determined  by  the  impedances  between  the 
wires  and  ground  along  the  line  and  at  central  offices  where  equipment  is  connected  to 
them.  The^voltage  to  ground  at  either  end  of  the  exposure  is  equal  to  the  product  of  the 
longitudinal  -current  and  the  impedance  to  ground  seen  looking  away  from  the  exposure 

at  that  end.  In  practice,  the 
variety  of  impedance  distri- 
butions encountered  is  almost 
infinite,  and  the  correspond- 
ing voltage  distributions  vary 
over  a  wide  range. 

If  voltage  to  ground  at  any 
point  where  protectors  are  lo- 
cated exceeds  the  operating 
voltage  of  the  protector,  the 
protector  operates  and  three 
things  happen: 

(a)  The  voltage  to  ground 
at  the  point  of  protector 
breakdown  is  reduced  to  a  low 
value.  This  causes  an  in- 
crease in  voltage  across  the 
protectors  at  the  opposite 
end,  and  in  most  cases  they 
also  will  operate. 

(6)  The  operation  of  the 
protectors  at  the  two  ends 
completes  a  loop  consisting  of 
the  telephone  circuit  and 

„,.,,.      «      ,  .      ,  ,-1        •,-        f  ^     •,    j.       ground,  so  that  the  induced 

Shielding  Resulting  from  the  Grounding  of  Conductors        -,,  -n  j_  ^ 

voltage  will  cause  current  to 


16        20       24       28       32       36 
NUMBER  OF  GROUNDED  WIRES 


of  an  Open-Tnre  Telephone  Line  (Courtesy  Bell  System) 


flow  through  both  protectors. 

(c)  The  voltages  to  ground  on  the  circuits  on  which  protectors  have  operated  are 
changed  and  redistributed,  and  the  voltages  on  the  other  telephone  circuits  are  also 
changed  and  redistributed,  owing  to  shielding. 

All  these  effects  take  place  within  a  very  short  time  after  the  induced  voltage  is  applied, 
so  that,  for  all  practical  purposes,  they  can  usually  be  considered  instantaneous. 

ACOUSTIC  DISTURBANCE.  Strictly  the  term  "acoustic  disturbance"  should  be 
used  only  with  reference  to  the  effect  of  an  abnormally  loud  sound  on  a  person  subjected 
to  it.  The  term  has  also  come  to  designate  a  noise  (usually  transient)  in  a  telephone 
receiver,  the  intensity  of  which  is  considerably  higher  than  that  of  speech.  It  is  produced 
by  an  excessive  voltage  across  the  terminals  of  the  receiver. 

Although  induced  voltages  usually  appear  in  equal  magnitudes  on  the  two  sides  of  a 
circuit,  the  protector  gaps  on  the  two  sides  of  the  circuit  discharge  in  an  unsymmetrical 
manner,  with  the  result  that  a  voltage  higher  than  normal  appears  across  the  circuit. 
When  this  occurs,  a  loud  noise  is  pro- 
duced in  the  receiver  of  a  telephone 
set  bridged  across  the  circuit,  causing 
acoustic  disturbance.  This  may  be 
produced  by  low-frequency  induction, 
lightning,  contacts  between  power 
and  telephone  circuits,  and  by  other 
causes. 

Figure  26  shows  oscillograph  traces 
of  voltages  measured  across  operating 
protector  blocks.  Each  outside  trace 


(a) 


Na.3j 

3  Element 
Oscillograph 

FIG.  26.     Oscillograms  Showing  Protector  Performance 
with  Impressed  Potentials  across  the  Protector  Blocks 


shows  the  voltage  across  its  related 

block  to  ground.    It  will  be  noted  that 

the  two  traces  are  not  identical.    The 

middle  trace  shows  the  resulting  voltage  across  the  circuit.    It  is  this  voltage  that  may 

cause  acoustic  disturbance.    The  very  jagged  outline  of  these  traces  indicates  that  many 

frequencies  are  present. 

PROBABILITY  FACTORS.  In  the  preceding  discussion,  a  number  of  factors  were 
mentioned  which  may  vary  between  different  occurrences  in  the  same  exposure.  Among 
them  axe  i 

(a)  The  impedance  in  the  faulted  circuit.     This  varies  with  the  location  of  faults, 


LOW-FKEQUENCT  INDUCTION  10-93 

effective  fault  resistance,  and  other  factors,  with  consequent  effect  on  the  magnitude  of 
the  fault  current. 

(6)  The  duration  of  the  fault  current.  This  varies  with  conditions  that  affect  the  speed 
of  operation  of  the  power-circuit  interrupting  devices. 

(c)  Longitudinal  voltage,  voltage  to  ground,  and  current  through  protectors.     These 
may  vary  widely  with  small  variations  in  the  locations  of  faults  when  they  occur  within 
the  exposure. 

(d)  Shielding  due  to  the  operation  of  protectors  on  telephone  circuits.    The  number  of 
protectors  that  operate  may  vary  widely,  depending  on  their  characteristics  and  the  mag- 
nitude of  the  induced  voltage. 

LOW-FREQUENCY  INDUCTION  CONTROL,  though  not  usually  required  under 
normal  operating  conditions,  must  be  planned  for  in  advance  where  service  may  be  affected 
by  abnormal  power-circuit  conditions  and  where  prompt  action  must  be  taken  to  restore 
normal  operating  conditions  when  abnormal  conditions  arise.  Cooperative  advance 
notices  of  construction  and  eoordinative  plans  are  essential  in  controiling  the  generally 
serious  effects  of  low-frequency  induction. 

The  low-frequency  coordination  of  power  and  communication  systems  may  be  accom- 
plished by  (1)  measures  in  the  power  system  to  limit  the  influence,  (2)  measures  in  the 
communication  system  to  limit  the  susceptiveness,  and  (3)  coordinated  location  of  lines 
or  other  means  to  reduce  coupling.  In  a  specific  situation,  one  measure  may  be  sufBcient 
or  two  or  more  measures-  may  be  required,  depending  on  the  conditions. 

POWER  SYSTEMS.  Measures  to  reduce  the  inductive  infiuence  of  power  systems 
should  be  directed  to  limiting  the  magnitude  of  unbalanced  currents  and  voltages,  par- 
ticularly under  abnormal  conditions,  and  to  reducing  the  duration  and  frequency  of  occur- 
rence of  faults.  Of  such  measures,  some  are  concerned  with  questions  of  Kne  and  system 
design  and  must  be  incorporated  in  the  construction  plans  and  others  may  be  added  later, 
if  found  necessary  as  the  result  of  operating  experience. 

Fault-resistive  Design  and  Construction.  Since  lightning  is  a  major  cause  of  faults 
on  power  systems,  the  developments  in  methods  of  lightning  protection  have  substantially 
aided  the  low-frequency  coordination  problem. 

Fault-current  Lmitation.  Resistors  or  reactors  in  the  neutrals^  of  power  systems  provide 
a  means  of  limiting  the  magnitude  of  the  residual  currents  except  when  double  faults  occur. 

Shielding.  Ground  wires  on  a  power  line,  though  they  may  increase  the  total  residual 
current,  provide  shielding  by  reducing  the  strength  of  the  external  electric  and  magnetic 
fields.  Under  fault  conditions,  ground  wires  generally  reduce  the  voltages  induced  in 
paralleling  communication  circuits.  The  effectiveness  of  such  shielding  depends  on  the 
impedance  of  the  shielding  conduetor  and  its  ground  connections.  Under  favorable  condi- 
tions, the  induced  voltage  at  60  cycles  may  be  reduced  about  40  per  cent  by  the  use  of 
one  wire,  and  about  60  per  cent  by  the  use  of  two  wires.  However,  such  shield  wires  may 
increase  the  normal  60-cycle  induced  voltage,  and  this  reaction  may  be  important  where 
ground-return  telegraph  circuits  are  involved. 

High-speed  Circuit  Breakers  and  Relays.  High-speed  relay  and  switching  systems 
have  been  developed  that  reduce  the  tune  duration  of  a  power-line  fault  to  the  order  of 
Vs  second  or  less  under  favorable  conditions,  thus  tencling  to  minimize  the  effects  of 
induction.  Because  of  the  expense,  high-speed  switching  can  seldom  be  justified  except 
on  important  high-voltage  transmission  systems. 

Improvement  in  Balance.  Low-frequency  induction  between  power  and  communica- 
tion lines  is  sometimes-  experienced  under  normal  operating  conditions.  On  grounded 
telegraph  and"  signal  lines,  the  trouble  usually  manifests  itself  by  a  chattering  of  telegraph 
instruments  or  by  false  signals.  Sometimes  power-line  transpositions  will  aid  in  these 
situations. 

COMMUNICATION  SYSTEMS.  la  general,  coordinative  measures,  applicable  to  the 
communication  system,  take  the  f orna  of  arrangements  or  devices  for  removing  or  counter- 
acting the  voltages,  the  voltage  to  ground,  or  the  current  through  telephone  protectors. 

Relay  Protectors.  The  short-circuiting  relay  (SCR)  protector  is  designed  for  applica- 
tion to  open- wire  telephone  Hnes  which  may  be  subjected  to  low-frequency  induction  of 
sufficient  magnitude  to  warrant  the  costs  of  providing  it.  This  device  employs,  for  each 
open-wire  circuit,  a  relay  which  short-circuits  the  usual  protector  blocks  associated  with 
the  circuit  and  grounds  the  circuit.  In  one  device  of  this  type  (Fig.  27),  a  master  relay 
controls  operation  of  all  of  the  individual  circuit  relay st  the  master  relay  operating  in  about 
0.01  second  and  the  individual  relays  about  0.15  second  later,  after  any  one  of  the  line 
protector  blocks  operates.  The  master  relay  operates  when  about  1  amp  or  more  of  60- 
cycle  current  flows  through  the  primary  winding  of  the  saturating  transformer  to  ground. 
This  produces  a  voltage  across  the  secondary  of  the  transformer,  operating  the  master 
"J"  type  relay,  causing  the  short-circuiting  relays  to  operate  and  short-circuit  each  line 


10-94 


TRANSMISSION   CIRCUITS 


and  its  associated  protector  blocks.  The  master  relay  can  be  adjusted  to  release  on  about 
0.8  amp  of  line  current.  Thus,  with  operation  of  the  pilot  relay,  all  relays  operate  on 
about  0.5  amp  direct  current  and  ground  all  wires  on  the  line  at  the  locations  where  this 
equipment  is  installed.  The  equipment  may  be  installed  at  any  outside  point  on  the  open- 
wire  line  or  in  the  central  office,  as  desired.  It  is  assembled  in  groups  for  application  to 
10  to  50  or  more  wires  and,  together  with  the  dry-cell  batteries  for  operating  the  circuit 
relays,  is  housed  in  standard  cable  terminal  boxes. 

Acoustic  Disturbance  Reducers.  One  of  the  most  effective  ways  of  reducing  acoustic 
disturbance  to  operators  is  to  shunt  their  receivers  with  a  device  that  will  have  a  high 
impedance  at  speech  level  and  a  low  impedance  at  acoustic-disturbance  level.  One  such 
device  that  has  proved  effective  consists  of  oppositely  poled  copper  oxide  rectifier  disks 
connected  across  the  receiver.  These  have  the  property  of  greatly  diminishing  impedance 
with  increasing  voltage. 

Shielding.  Shielding  on  a  telephone  line  may  be  effected  by  special  grounded  conduc- 
tors, by  working  conductors,  or  by  cable  sheaths.  Miscellaneous  structures,  such  as  pipe 


TO  LINE         J 

TOUNE 

PROTECTOR 

PROTECTOR 

ATIN6 

B 

LOCKS 

B 

LOCK 

5 

31  L 
RATING 
FORMER) 

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-00 

f 

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TO  ALL   UNITS 
IN    PROTECTOR 

jr                        £  ! 

1  ... 

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X 

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P 

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FIG.  27.    A-c  Type  Short-circuiting  Relay  Protector  Circuit — Multi-grounding  (Report  41,  J.C.P.C.) 

lines  or  rails  in  the  immediate  vicinity  of  an  exposure,  also  introduce  shielding.  A  high- 
conductivity  shield  wire,  well  grounded  at  the  ends  of  the  exposure  and  at  intermediate 
points,  may  reduce  the  induced  voltage  by  as  much  as  40  per  cent  at  60  cycles. 

The  lead  sheath  of  a  2  5/g-m.-diameter  aerial  telephone  cable,  if  effectively  grounded 
at  the  ends,  reduces  the  voltage  induced  in  the  conductors  within  the  cable  by  about  50 
per  cent  at  6X3  cycles.  Ihe  shielding  secured  from  the  sheath  of  an  underground  cable  of 
this  size  is  also  about  50  per  cent.  The  large  number  of  conductors  in  a  cable  affords  mutual 
shielding  which  varies  from  a  negligible  to  a  considerable  amount  (sometimes  exceeding 
95  per  cent),  depending  upon  many  factors,  important  among  which  is  the  extent  of  the 
cable  beyond  the  ends  of  the  exposure  and  the  grounding  conditions  on  the  circuits  at  their 
terminals.  If  two  or  more  cables  are  close  together  through  an  exposure,  each  benefits  by 
the  shielding  action  of  the  others,  so  that  the  shielding  increases  with  the  number  of  cables. 

If  the  lead  sheath  of  the  cable  is  surrounded  by  magnetic  material,  as  by  armoring  or 
placing  cable  in  iron  pipe,  the  shielding  may  be  largely  increased.  With  a  form  of  iron 
tape  armored  cable,  shielding  at  60  cycles  may  be  SO  per  cent  or  more,  assuming  effective 
grounding. 

Other  Measures.  Sometimes  drainage  and  neutralizing  transformers  may  be  of  use. 
Drainage  is  achieved  by  grounding  the  midpoint  of  a  coil  connected  between  the  two  sides 
of  the  communication  circuit,  the  coil  being  wound  in  such  a  way  that  it  presents  a  low 
impedance  to  longitudinal  currents  and  a  high  impedance  to  alternating  currents  in  the 
metallic  circuit.  The  voltage  to  ground  on  the  communication  circuit  at  the  point  where 
the  drainage  is  connected  is  limited  to  the  voltage  drop  over  the  impedance  of  the  coil 
and  ground  connection. 

The  neutralizing  transformer  introduces  into  the  exposed  communication  wires  a  voltage 
in  opposition  to  the  disturbing  voltage,  thereby  partly  neutralizing  it,  as  described  under 
"Noise  Induction  Mitigation'*  in  article  32. 

On  account  of  the  cost  and  the  operating  limitations  that  these  measures  impose  on 
the  use  of  carrier,  d-c  telegraph,  and  testing,  they  have  only  occasionally  been  employed 
in  the  commercial  telephone  plant.  Neutralizing  transformers  have,  however,  been  used 
on  telegraph  circuits  and  for  special  protection  purposes  on  other  types  of  communication 
and  signal  circuits. 


BIBLIOGRAPHY  10-95 


BIBLIOGRAPHY 

Principles  and  Practices 

1.  Reports  of  Joint  General  Committee  of  Edison  Electric  Institute  and  Bell  Telephone  System: 

(a)  Principles  and  Practices  for  the  Inductive  Coordination  of  Communication  Systems,  Dec.  9,  1922. 

(6)  Principles  and  Practices  for  the  Joint  Use  of  Wood  Poles  by  Supply  and  Communication  Com- 
panies, Feb.  15,  1926. 

(c)  Inductive  Coordination — Allocation  of  Costs  between  Supply  and  Communication  Companies, 
Oct.  15,  1926. 

2.  Reports  of  Joint  General  Committee  of  American  Railway  Association  (now  Association  of  Amer- 

ican Railroads)  and  Bell  Telephone  System: 

Inductive  Coordination  of  Railway  Electrical  Supply  Facilities  and  the  Communication  Facilities  of 
the  Bell  System,  Apr.  15,  1931,  and  Sept.  1,  1932. 

3.  Reports  of  Joint  General  Committee  of  the  Edison  Electric  Institute  and  the  Western  Union  Tele- 

graph Company: 
Physical  and  Inductive  Coordination  of  Electrical  Supply  and  Communication  Systems,  July  22,  1935. 

4.  Reports  of  the  Joint  General  Committee  of  the  Edison  Electric  Institute  and  Bell  Telephone  System 

entitled  Physical  Relations  between  Electrical  Supply  and  Communication  Systems  (EJ3.I.  Pub. 
M5). 

5.  Year  Book  of  the  American  Standards  Association  (ASA). 

6.  Report  of  the  Joint  General  Committee  on  Plant  Coordination  of  the  Edison  Electric  Institute 

and  Bell  Telephone  System  entitled  Joint  Pole  Practices  for  Supply  and  Communication  Circuit* 
(E.E.I.  Pub.  M12),  Oct.  29,  1945. 

7.  Safety  Rules  for  the  Installation  and  Maintenance  of  Electric  Supply  and  Communication  Lines 

(National  Bureau  of  Standards  Handbook  E32,  Sept.  23,  1941)   (N.E.S.  Code — 5th  Edition). 

8.  Discussion  of  the  National  Electrical  Safety  Code,  Fifth  Edition,  Part  2,  and  Grounding  Rules 

(National  Bureau  of  Standards  Handbook  H39),  July  15,  1944. 

9.  Specification  1-B-l  for  Communication  Lines  Crossing  the  Tracks  of  Railroads,  Association  of  Amer- 

ican Railroads. 

Tecnnical  Data 

1.  Bibliography  on  Inductive  Coordination  by  American  Committee  on  Inductive  Coordination  (1925). 

(List  of  articles  and  reports  published  before  Jan.  1,  1925.) 

2.  Symposium  on  Coordination  of  Power  and  Telephone  Slant,  AJ.E.E.  Trans.,  Vol.  50,  437-478. 

Part  I:  Trends  in  Telephone  and  Power  Practice  as  Affecting  Coordination,  by  W.  H.  Harrison 
and  A.  E.  Silver.  Part  II:  Status  of  Joint  Development  and  Research  on  Noise  Frequency  In- 
duction, by  H.  L.  Wills  and  0.  B.  BlackwelL  Part  III:  Status  of  Joint  Development  and  Research 
on  Low-frequency  Induction,  by  R.  N.  Con  well  and  H.  S.  Warren.  Part  IV:  Status  of  Coopera- 
tive Work  on  Joint  Use  of  Poles,  by  J.  C.  Martin  and  H.  L.  Huber.  (These  papers  include 
references  to  the  more  important  papers  published  before  January  1931.) 

3.  Experimental  Studies  of  Arcing  Faults  on  a  75-kv  Transmission  Line,  AJ.E.E.  Trans.,  VoL  50, 

1469. 

4.  Effects  of  Rectifiers  on  System  Wave  Shape,  AJ.E.E.  Trans.,  Vol.  53,  54* 

5.  Petersen  Coil  Tests  on  140-kv.  System,  AJ.E.E.  Trans.,  Vol.  53,  63. 

6.  Iron  Shielding  for  Telephone  Cables,  AJ.E.E.  Trans.,  VoL  53,  274. 

7.  Overvoltages  on  Transmission  Lines,  AJ.E,E.  Trans.,  Vol.  53,  1301. 

8.  Joint  Use  of  Poles  with  6900-volt  Lines,  AJ.E.E.  Trans.,  VoL  52,  890. 

9.  Relay  Handbook  and  Supplement,  N.E.L.A.,  1931. 

10.  N.E.L.A.  Publication  118,  Technical  Theory  of  Inductive  Coordination. 

11.  N.E.L.A.  Publication  144,  Some  Features  of  Telephone  and  Telegraph  Systems. 

12.  N.E.L.A.  Publication  147,  Cable  Circuit  Noise  and  Power  Distribution  Systems. 

13.  N.E.L.A.  Publication  153,  The  Effects  on  Noise  Frequency  Induction  of  Grounded  Neutral  Generators. 

14.  N.E.L.A.  Publication  212,  Low-frequency  Induction. 

15.  NJ2.L.A.  Publication  239,  Generator  Wave  Shape. 

16.  E.E.L  Publication  A-13,  Power  System  Wave  Shape. 

17.  Low-frequency  Power  Induction  and  Its  Effect  on  D-c  Telegraph  Operation,  presented  at  annual 

meeting,  Telegraph  and  Telephone  Section,  A.R.A.,  June  12r  1934. 

18.  Demonstration  of  Low-frequency  Induction  between  Power  and  Telephone  Circuits,  also  presented 

at  above  A.R.A.  meeting. 

19.  Cooperation  is  Kevnote  in  Rectifier  Supply  Coordination,  by  F.  E.  Sanford  and  V.  G.  Rettig, 

Edison  Elec.  Inst.  Bull,  August  1935. 

20.  Earth  Resistivity  and  Geological  Structure,  AJ.E.E.  Trans.,  Vol.  54,  1153. 

21.  Measurement  of  Telephone  Noise  and  Power  Wave  Shape,  AJ.E.E.  Trans.,  Vol.  54,  1307. 

22.  Engineering  Reports  of  the  Joint  Subcommittees  of  the  Edison  Electric  Institute  and  the  Bell 

Telephone  System: 

Vol.  II,  Report  of  Proj.  Comm.  No.  10,  April  1932. 
VoL  II,  Report  12,  April  1932. 
VoL  II,  Report  14,  April  1932. 
Vol.  IV,  Report  34,  January  1937. 
Vol.  IV,  Report  36,  January  1937. 
VoL  IV,  Report  38,  January  1937. 
VoL  V,  Report  41,  January  1943. 
VoL  V,  Report  44,  January  1943. 
VoL  V,  Report  45,  January  1943. 
Vol.  V,  Report  46,  January  1943. 
VoL  V,  Report  48,  January  1943. 

23.  A.I.E.E.  Committee  Report,  AJ.E.E.  Trans.,  VoL  65,  417. 

24.  Provisional  Report  24  (Technical  Report  2G-2),  A  Study  of  Pole  Strength  in  Jointly  Used  Poles,  of 

the  Joint  Subcommittee  on  Development  and  Research  of  the  E.E.I,  and  B.T.S.,  Aug.  22,  1938. 

25.  AJ.E.E.  Technical  Report  47-82,  Joint  Use  of  Pole  Lines  for  Rural  Power  and  Telephone  Sermcest 

by  J.  W.  Campbell,  L.  W.  Hill,  L.  M.  Moore,  and  H.  J.  Scholz,  December  1946. 


SECTION  11 
ELECTRICAL  MEASUREMENTS 


FREQUENCY  MEASUREMENTS 
ART>  BY  WARREN  A.  HARRISON  PAGE 

1.  Definitions 02 

2.  Using  a  Time  Standard 04 

3.  Using  a  Frequency  Standard 05 

4.  Employing  Circuit  Element  Selectivity  11 

5.  Electromagnetic  Phenomena 14 

MEASUREMENT  OF  PRIMARY 
ELECTRICAL  QUANTITIES 

BY  J.  G.  FERGUSON 

6.  Measurement  of  Current 16 

7.  Measurement  of  Voltage 17 

8.  Resistance  Standards 18 

9.  Capacitance  Standards 20 

10.  Inductance  Standards 22 

11.  Measurement  of  Resistance 23 

12.  Measurement  of  Capacitance  and  Con- 

ductance    24 

13.  Measurement  of  Inductance  and  Effec- 

tive Resistance 27 

14.  Signal  Generators  and  Detectors 30 

WIRE  LINE  MEASUREMENT 
BY  H.  J.  FISHER 

15.  Transmission  Measurements 32 

16.  Noise  Measurements 36 

17.  Cross-talk  Measurement 37 

18.  Echo  Testing  of  Lines 38 

19.  Square  Wave  Testing 39 

20.  A-c  Bridge  Method  of  Locating  Irregu- 

larities in  Line  Impedance 39 

21.  D-c  and  Low-frequency  line  Testing. . .  41 


ROUTINE  MEASUREMENTS  ON  A-M 
AND  F-M  BROADCAST  RECEIVERS 

ART.  -^T  ^*    O-   SwiXYARD  PAGE 

22.  Overall  A-m  Receiver  Measurements.  .  .     43 

23.  Single-stage  Measurements 48 

24.  Miscellaneous    Measurements    on    A-m 

Receivers 50 

25.  F-m  Receiver  Measurements 50 

26.  Overall  Performance  Tests 51 

27.  Single-stage  Measurements 52 

WAVE  ANALYSIS 
BY  E.  PETERSON 

28.  Wave  Characteristics 54 

29.  General  Analyzer  Requirements 56 

30.  Methods  of  Wave  Analysis 58 

31.  Spectrographs 65 

MICROWAVE  MEASUREMENTS 

BY  E.   W.    HotTQHTON 

32.  Impedance  Measurements 70 

33.  Absolute  Power  Measurements 77 

34.  Attenuation  Measurements 82 

35.  Frequency  Measurements 84 

SIGNAL  GENERATORS  AND  POWER 
MEASUREMENT 
BY  F.  J.  GAFFNEY 

36.  Oscillators  for  Signal  Generator  Use 89 

37.  Modulation  of  Signal  Generators 93 

38.  Standardization  of  Output  Power 96 

39.  Attenuator  Design 99 

40.  Shielding  Problems 101 

41.  Power  Measurement 102 


11-01 


ELECTRICAL  MEASUREMENTS 
FREQUENCY  MEASUREMENTS 

By  Warren  A.  Marrison 

1.  DEFINITIONS 

By  frequency  is  meant  the  number  of  times  any  periodic  phenomenon  recurs  in  a  standard 
unit  of  time.  In  general  it  is  stated  as^the  number  of  complete  vibrations,  or  oscillations, 
or  revolutions  performed  in  1  sec. 

Occasionally  other  units  of  time  are  used,  but  usually  they  are  specified  so  there  is  little 
or  no  ambiguity.  Thus,  we  speak  of  revolutions  per  minute.  Sometimes  tuning  forks 
are  stamped  DV  or  VS  after  the  number  designating  the  frequency.  DV,  which  stands 
for  "double  vibrations,"  means  frequency  as  denned  above,  that  is,  the  number  of  com- 
plete periods  per  second,  VS,  which  stands  for  "vibrations  per  second,"  designates  twice 
the  true  frequency. 

The  period  of  a  cyclic  or  periodic  phenomenon  is  the  time  duration  of  one  complete 
cycle.  The  rates  of  slow  periodic  phenomena  are  usually  expressed  in  terms  of  the  period. 
For  example,  the  period  of  a  "seconds"  pendulum  is  2  sec. 

Frequency  may  also  be  designated  as  angular  velocity,  especially  in  expressions  repre- 
senting the  rate  as  a  trigonometric  function  of  time.  Thus  in  expressions  of  the  type 

I  =  IQ  sin  (coi  +  a) 
the  constant,  «  =  2x/,  may  be  considered  as  an  angular  velocity. 

Frequency  may  be  expressed  in  terms  of  wavelength  when  the  velocity  of  wave  propaga- 
tion in  a  medium  is  assumed.  This  is  most  often  used  in  discussions  of  electromagnetic 
radiation  where  the  velocity  is  approximately  that  of  light,  c  =  2.998  X  1010  cm  per  sec. 
The  general  relationship  between  frequency,  velocity,  and  wavelength  is 

Velocity          v 
f  (OT  ">  =  Wavelength  =  X 

A  standard  of  frequency  differs  from  most  other  working  standards  in  that  it  is  a  rate 
and  cannot  be  represented  completely  by  a  physical  body  that  can  be  preserved.  Although 
the  frequency  of  a  bar,  or  other  simple  shape,  can  be  defined  approximately  for  a  given 
mode  of  vibration  in  terms  of  its  dimensions,  density,  elasticity,  and  coupling  to  other 
bodies,  the  effects  of  these  factors  will  not  remain  constant  with  as  great  accuracy  as  that 
to  which  the  resultant  frequency  can  be  measured  in  terms  of  astronomical  time,  that  is, 
in  terms  of  the  rate  of  the  earth's  rotation  on  its  axis. 

In  the  present  system  of  measurements,  the  primary  standard  of  frequency  is  the  rate 
of  the  earth's  rotation,  since  all  measurements  of  frequency  are  referred  directly  or  in- 
directly to  the  second  of  the  cgs  system  which  is  by  definition  1/86,400  of  a  mean  solar  day. 

By  the  application  of  recently  discovered  atomic  or  molecular  resonance  phenomena  to 
the  measurement  of  frequency  and  time  it  may  be  found  possible  eventually  to  place  all 
such  measurements  on  a  more  nearly  absolute  basis,  quite  independent  of  the  stability 
of  masses  of  matter  such  as  now  comprise  the  control  elements  in  reference  standards  of 
frequency  and  time,  and  independent  of  variations  known  to  exist  in  astronomical  time 
itself.  Some  of  these  atomic  and  molecular  resonance  phenomena  are  in  the  range  for 
which  continuous  oscillations  at  ultra-high  frequency  may  be  produced  by  modern  vacuum- 
tube  means,  indicating  the  possibility  of  making  frequency  comparisons  of  high  precision 
throughout  the  entire  range  of  continuous  magnetic  waves.  The  possibility  of  so  using 
these  resonance  phenomena  was  discussed  by  Professor  I.  I.  Rabi  in  a  talk  before  the 
American  Physical  Society  and  the  American  Association  of  Physics  Teachers  in  January, 
1945. 

The  great  importance  of  this  impending  development  lies  in  the  idea  that  the  atomic 
and  molecular  resonance  phenomena,  being  properties  of  independent  elementary  particles 
of  matter,  appear  to  be  in  certain  cases  completely  independent  of  temperature,  pressure,, 
and  other  ambient  conditions,  which  in  various  degrees  affect  the  behavior  of  all  existing 
practical  standards  of  frequency  and  time. 

11-02 


DEFINITIONS 


11-03 


Table  1.     Some  Useful  Frequency  Formulas 


1.  A-c  generator  with  alternate  N  and  S  poles: 
Inductor  generator: 

2.  Electrical  resonant  circuit  with  L  and  C: 

Same  with  L,  C,  and  R  in  series: 

3.  Frequency  of  electromagnetic  radiation  (X  =  wavelength) : 


/  = 


No.  of  poles  X  rps 


/  =  No.  of  poles  X  rps 


3.00  X  1010 
X(cm) 


Frequency  of  sound  vibrations  in  any  medium:  / 

4.  If  a  condenser  is  charged  to  voltage  E  and  completely  discharged 
/tunes  a  second  into  a  current-measuring  device:  / 


_  1      /Y 
X    V 


'oung's  modulus 


Density 

Mean  current 
C  X  E 


5.  The  period  T  of  a  simple  pendulum  with  double  amplitude  =  29 :    T  =  2*"\  /-  (  1  -f     sin2  -  •  •  •  I 

\*\         4         2         / 

The  frequency  for  small  amplitude:  /  =  —  -V/l 

Conical  pendulum  (angle  B  to  vertical) : 
Vertical  pendulum  (mass  supported  on  spring) : 

Torsion  pendulum: 

where  rj, 1,  and  R  are  the  torsion  modulus,  length,  and  radius  of 
the  supporting  member,  and  I  is  the  moment  of  inertia,  of  the 


6.  Stretched  string: 
Overtones  are  2/,  3/,  etc. 

7.  Uniform  rod,  free-free,  longitudinal: 
Overtones  are  2/,  3/,  etc. 

8.  Uniform  round  rod,  free-free,  torsion: 
Overtones  are  2f,  3/,  etc. 

9.  Long  air  column  open  at  both  ends: 
7  =  ratio  of  the  specific  heats. 
Overtones  are  2/,  3/,  etc. 

Long  air  column  open  at  one  end: 
Overtones  are  3/,  5/,  etc. 

10.  Straight  free-free  bar  in  flexure: 

k  is  radius  of  gyration  of  section. 
Overtones  are  (5/3)  2/,  (7/3)  2f,  (9/3)  2/,  etc. 


Density 


For  flexural  vibrations  in  generalt  tuning  forks,  reeds,  etc.,  having    /  =»  K  —^  A/— 5S*  s  m        us 


uniform  section: 
where  K  is  a  constant  for  a  given  shape  and  mode  of  vibra- 
tion. 

11.  For  any  note  in  the  equally  tempered  scale  where  n  is  the  total    f±n 
number  of  semitones  above   (+)   or  below  (—  )  middle  C. 
(A  =  440  cps.) 

Exact  frequency  ratios  between  successive  notes  in  the  major 
diatonic  scale  are:  9/8,  10/9,  16/15,  9/8,  10/9,  9/8,  16/15. 


Density 


220 


12.  Where  I  is  the  distance  between  nodes  on  Lecher  wires  or  in 
a  coaxial  conductor: 


_._  3  X  1010 
2Z 


11-04 


ELECTRICAL  MEASUREMENTS 


The  most  accurate  reference  standards  of  frequency  in  use  at  present  are  vacuum-tube 
oscillators  whose  frequencies  are  controlled  by  mechanical  vibrators  made  of  quartz 
crystal.  A  special  advantage  of  the  oscillator  type  of  standard  is  that  its  output  of  con- 
stant-frequency current  can  be  sent  over  suitable  communication  channels  and  used 
anywhere  as  a  reference  standard  of  frequency.  Such  standards  are  maintained  contin- 
uously by  the  laboratories  of  the  National  Bureau  of  Standards,  the  Bell  System,  and 
many  others  in  America  and  abroad. 

Oscillators  of  somewhat  less  accuracy  have  been  built  employing  tuning  forks  or  bars 
of  metal  for  the  frequency-controlling  element.  Usually  these  are  coupled  to  the  vacuum- 
tube  circuit  through  electromagnetic  means,  but  many  such  oscillators  have  been  built 
using  magnetostriction  and  electrostatic  attraction  for  the  coupling  means.  Prior  to  the 
general  use  of  quartz  for  precise  control,  such  oscillators  were  used  in  most  frequency 
standard  installations.  Now  they  are  used  chiefly  where  the  advantage  of  direct  low- 
frequency  control  outweighs  the  requirement  for  extreme  accuracy. 

Although,  for  the  greatest  accuracy,  frequency  is  defined  and  measured  in  terms  of 
astronomical  time,  preferably  by  means  of  a  combined  time  and  frequency  reference  stand- 
ard, there  are  many  cases  in  which  a  good  estimate  of  its  value  can  be  made  from  a  knowl- 
edge of  means  controlling  it  or  responding  to  it.  Table  1  contains  a  number  of  formulas 
that  may  be  used  for  the  approximate  determination  of  frequency  under  a  variety  of  con- 
ditions. 

Frequency  measurements  find  their  chief  applications  in  electrical  communication  where 
they  are  used  in  the  study  and  adjustment  of  various  oscillators  and  electrical  networks 
such  as  niters  and  equalizers.  This  applies  in  varying  importance  from  the  lowest  frequen- 
cies employed  in  d-c  telegraph  to  the  highest  used  in  ultra-short-wave  radio. 

They  are  involved  in  the  control  of  power  frequencies  and  in  certain  time  systems.  They 
are  used  in  measuring  linear  and  angular  velocity  and  acceleration,  for  the  study  of  vibra- 
tion in  mechanical  systems,  and  for  the  most  precise  determinations  of  electrical-circuit 
constants. 

They  are  of  ever-increasing  importance  in  basic  physical  studies  involving  relations 
between  astronomical  time,  the  velocity  of  light,  and  resonance  phenomena  in  atoms  and 
molecules. 

2.  USING  A  TIME  STANDARD 

COUNTING.  Since  frequency  is  defined  as  the  number  of  recurrences  of  a  cyclic  phe- 
nomenon per  unit  of  time,  the  most  direct  method  of  its  measurement,  and  at  the  same 
time  the  most  precise,  is  to  count  the  total  number  of  cycles  during  a  known  time  interval 
and  to  divide  by  the  number  of  elapsed  seconds.  The  only  inaccuracies  in  this  method 
are  in  determining  the  time  interval  and  in  estimating  the  number  of  cycles.  If  the  time 
error  can  be  assumed  to  be  negligible,  and  if  the  frequency  is  constant,  the  accuracy  may 
be  increased  to  any  extent  by  increasing  the  duration  of  the  measurement. 

Depending  on  the  nature  and  the  frequency  of  the  phenomena  to  be  measured,  and  the 
accuracy  required,  the  instrumentation  may  vary  over  a  very  wide  range.  For  low  fre- 
quencies, and  with  relatively  low  accuracy,  a  stop  watch  may  be  used  to  count  recurrences 
over  intervals  of  a  few  seconds.  For  greater  accuracy,  and  for  frequencies  too  high  for 
direct  perception,  some  automatic  registering  or  totalizing  means  is  required.  By  means 

of  a  chronograph  or  oscillograph,  a  recording 
can  be  made  of  the  recurrences  during  a  speci- 
fied time  interval  which  may  be  analyzed  at 
leisure.  By  various  counting  means  a  continu- 
ous total  may  be  produced  from  which  the  fre- 
quency may  be  determined  with  great  accuracy 
in  any  desired  time  interval. 

SYNCHRONOUS  CLOCK.  This  in  effect  is 
the  method  used  in  calibrating  the  most  accu- 
rate frequency  standards  of  the  quartz-con- 
trolled oscillator  type.  The  actual  method  con- 
sists in  operating  a  synchronous  clock  from  the 
standard  frequency  source  and  in  measuring 
the  rate  of  the  clock  in  terms  of  observatory 
time  signals  (astronomical  time)  by  means  of  a 
chronograph.  Since  the  accuracy  of  individual 


Fi<3.  1.     Submultiple  Generator  of  the  Miilti- 

vibrator  Type.     Capacitors  C  are  for  coupling 

only. 


time  signals  may  be  somewhat  better  than  0.01  sec,  the  day-by-day  accuracy  of  this 
method  is  of  the  order  of  1  part  in  ten  million.  When  the  constancy  of  an  oscillator 
justifies  the  use  of  a  longer  interval  between  checks,  the  accuracy  of  determination  may 


USING  A  FREQUENCY  STANDARD 


11-05 


be  increased  correspondingly.  In  this  manner  the  rates  of  the  best  crystal  oscillators 
may  be  determined  in  terms  of  astronomical  time  with  an  accuracy  better  than  1  part 
in  a  hundred  million. 

When  accuracies  of  this  order  are  involved,  the  variations  in  astronomical  time  itself 
should  be  taken  into  account,  since  changes  in  the  rate  of  the  earth  in  excess  of  1  part  in  a 
hundred  million  have  been  observed  from  time  to  time.  The  largest  such  variation  in 
recent  years  occurred  in  1918  and  amounted  to  about  1  part  in  thirty  million. 

Since  the  frequency  of  a  crystal  oscillator,  best  suited  for  use  as  a  precise  frequency 
standard,  is  too  high  to  operate  a  synchronous  motor  directly,  a  circuit  known  as  a  sub- 
multiple  generator  is  used  to  _  ^ 

produce  a  frequency  which  is  .  >""'  | 1  /and 

an  exact  submultiple  of  the 
original.  Two  or  more  stages 
may  be  required,  depending 
on  the  ratio  of  the  end  fre- 
quencies. In  one  system  that 
has  operated  continuously  for 
over  10  years,  the  ratios  5  X 


Output 


FIG.  2.     Submultiple  Generator  of  the  Regenerative  Modulator 
Type 


5X4  are  used  to  operate  1000-cycle  motors  from  a  100,000-eycIe  primary  standard.  Simi- 
lar means  may  be  used  to  obtain  frequencies  in  any  range  convenient  for  distribution  or  for 
measurements  which  bear  any  exact  rational  relation  to  the  primary  control  frequency. 
Two  of  the  more  important  means  for  frequency  subdivision  are  illustrated  in  Figs.  1  and  2. 
Although  the  former  is  most  widely  used,  the  latter  has  the  advantage  that  no  output  is 
produced  in  the  absence  of  an  input. 


3.  USING  A  FREQUENCY  STANDARD 
(Frequency  Comparison) 

Methods  for  frequency  comparison  may  be  divided  broadly  into  two  main  classes: 
those  in  which  the  actual  number  of  cycles  difference  per  unit  of  time  may  be  determined 

between  the  unknown  frequency  and  some  simple 
exact  fraction  or  multiple  of  the  standard,  and  those 
in  which  some  approximation  is  used  that  does  not 
permit  of  actual  counting.  The  first  includes  the  va- 
rious beat  methods,  the  accuracy  of  which  is  limited 
only  by  the  stability  of  the  sources  and  the  duration 
of  observations.  An  accuracy  of  comparison  of  1  part 
in  1010  may  be  obtained  by  some  of  these  methods 
under  good  conditions.  The  latter  includes  (1)  meth- 
ods in  which,  for  convenience  and  speed  of  operation, 
a  calibrated  interpolation  oscillator  is  used  to  interpo- 
late between  standard  frequency  values,  (2)  methods 
FIG.  3.  Balanced  Vacuum  Tube  Mod-  in  which  circuit  selectivity,  or  other  non-synchronous 
ulator  for  Observing  Low-frequency  means,  are  used  to  indicate  frequency  in  some  part  of 
a  the  system,  and  (3)  methods  in  which  the  frequency 

of  the  source  is  in  such  a  high  range  that  the  low-frequency  beat  contains  irregularities 
and  cannot  be  treated  as  a  sine  wave  and  used  in  any  direct  counting  procedure.  The 
accuracy  of  measurement  in  the  last  main  class  rarely  ex- 
ceeds 1  part  in  a  million  and  may  vary  over  a  wide  range, 
depending  on  the  particular  apparatus  used  and  the  skill  of 
the  observer. 

ZERO  BEAT.  The  simplest,  most  direct,  and  most  accu- 
rate methods  are  those  for  comparing  two  frequencies  which 
have  nearly  the  same  value.  If  the  two  sources  are  directed 
into  a  modulator  so  that  the  low-frequency  second-order 
modulation  produced  goes  through  a  d-c  meter,  the  meter 
reading  will  vary  periodically  at  a  rate  which  is  the  difference 
between  the  two  input  frequencies.  Thus  the  beats  may  be 
counted  over  a  suitable  interval  and  the  number  per  second 
thus  determined  and  added  to  or  subtracted  from  the  stand- 
ard to  obtain  the  unknown  value.  If  the  two  frequencies  are 
alike,  there  is  no  response,  hence  the  term  "zero  beat."  Figures  3  and  4  illustrate  typical 
-modulator  circuits  useful  for  this  method. 


FIG.  4.  Copper  Oxide  Modu- 
lator for  Observing  Low-fre- 
quency Beats 


11-06 


ELECTRICAL  3MEASUREMENTS 


If  a  third-order  modulator  is  used,  such  as  silicon  carbide,  the  greatest  response  is  ob- 
tained -with  input  frequencies  near  the  ratio  of  2  :  1.  If  higher-order  modulation  products 
are  used,  as  by  overloading  vacuum  tubes  or  other  modulators,  appreciable  response  can 
be  obtained  even  when  the  input  frequencies  are  related  as  m  :  n,  provided  that  the  product 
of  the  integers  m  and  n  is  not  too  great.  In  this  case  beats  are  obtained  between  the  nth 
harmonic  of  one  input  and  the  rath  harmonic  of  the  other.  The  numerical  accuracy  ob- 
tained is  proportional  to  the  ratio  of  the  actual  beat  frequency  observed  to  the  frequency 
of  the  particular  harmonic  involved  in  the  measurement.  If  the  beat  frequency  obtained 
in  this  way  is  too  high  to  observe  directly,  other  means  may  be  used  to  determine  it. 

The  accuracy  obtainable  by  this  general  method  may  be  very  high.  For  example,  if 
the  two  frequencies,  being  in  the  ratio  of  nearly  1:1,  are  about  100,000  cycles,  and  if  a 
beat  of  about  1  cycle  in  10  seconds  is  obtained,  an  accuracy  of  only  10  per  cent  in  the 
observed  beat  frequency  corresponds  to  an  accuracy  of  1  part  in  ten  million  in  the  compar- 
ison of  the  two  high  frequencies.  Highly  stable  sources  may  be  compared  readily  by  this 
means  with  an  accuracy  exceeding  1  part  in  1010. 

OTHER  BEAT  METHODS.  If  the  beat  frequency  obtained  from  a  modulator  is  too 
high  to  observe  directly,  it  may  be  measured  by  any  other  method  suitable  for  the  par- 
ticular range  encountered.  For  continuous  comparisons  the  most  precise  method  is  to 
operate  a  synchronous  clock  from  the  beat  frequency  and  to  compute  the  frequency  from 
its  rate.  A  small  percentage  change  in  the  rate  of  the  undetermined  high  frequency  will 
cause  a  much  larger  percentage  change  in  the  rate  of  the  clock.  For  example,  if  the  stand- 
ard were  100,000  cycles  and  the  undetermined  frequency  in  the  neighborhood  of  100,100 
cycles,  the  multiplying  factor  would  be  1000. 

If  merely  an  indication  of  moderate  accuracy  is  required,  without  integrating  or  record- 
ing, a  commercial  frequency  meter,  such  as  the  vibrating-reed  type,  can  be  used  giving  a 
direct  reading  accurate  to  1  cycle  or  a  little  better.  The  beat  frequency  must  of  course 
fall  in  the  range  of  the  particular  instrument  employed. 

A  very  satisfactory  direct-reading  means  for  measuring  the  beat  frequency  in  the  range 
from  one  cycle  to  about  200  cycles  per  second  is  illustrated  in  Fig.  5.  The  modulator 

output  operates  a  relay  having  a  transfer  con- 
tact combination.  A  condenser  with  capac- 
itance C  is  charged  to  voltage  E  at  the  beat 
frequency  /  times  per  second.  The  meter 
then  reads  /  X  C  X  E,  which  is  directly  pro- 
portional to  /.  The  factors  C  and  E  may  be 
chosen  in  relation  to  the  sensitivity  of  the 
meter  so  as  to  obtain  a  satisfactory  scale 
reading.  For  very  low  frequencies,  either  a 
ballistic  type  of  meter  should  be  used,  or  a 
filter  should  be  included  in  the  meter  circuit, 
in  order  to  reduce  meter  fluctuation.  Greater 
FIG.  5.  Direct-reading  Beat  Frequency  Indicator  accuracy  may  be  obtained,  at  the  expense  of 

the  direct-reading  feature,  by  balancing  the 

voltage  across  a  resistor,  used  instead  of  the  meter,  against  a  fraction  of  the  voltage  E,  thus 
producing  a  null  instrument.  Both  methods  are  suitable  for  continuous  recording  with 
commercial  sensitive  d-c  recorders  and  are  capable  of  high  overall  accuracy. 

Direct-reading  instruments  on  the  above  principle  have  been  built  in  which  the  relay 
is  replaced  by  electron  trigger  tubes  in  order  to  extend  the  usable  range  of  the  beat  fre- 
quency. 

If  the  beat  frequency  is  too  high  to  measure  accurately  by  any  direct  method,  it  can  be 
determined  in  terms  of  a  lower  standard  frequency  by  using  a  second  modulating  stage. 
This  process  can  be  carried  out  through  a  number  of  stages  if  desired  so  that,  regardless  of 
the  value  of  the  original  undetermined  frequency,  the  final  beat  is  low  enough  to  be  meas- 
ured accurately  by  some  direct  method.  In  one  method  that  has  been  worked  out  in 
practice  for  measuring  high  frequencies,  the  frequency  standard  is  made  available  in 
multiples  of  1,000,000  cycles,  100,000  cycles,  10,000  cycles,  etc.,  so  that,  by  successive 
stages  of  modulation,  a  frequency  determination  is  made  in  the  range  between  5000  and 
30,000  kc  in  terms  of  a  low-frequency  beat  and  exact  multiples  of  the  several  decade 
standards. 

The  frequency  range  over  which  this  principle  may  be  applied  is  limited  only  by  the 
stability  of  the  high-frequency  oscillations.  So  long  as  the  final  beat  frequency  can  be 
measured  as  a  continuous  sine  wave,  the  limiting  accuracy  is  about  the  same  as  for  the 
zero-beat  method.  When,  as  is  true  at  present  of  many  ultra-high-frequency  oscillators, 
the  final  beat  signal  contains  such  random  variations  that  the  cycles  cannot  be  actually 
counted,  some  uncertainty  enters  even  the  best  determinations  of  frequency.  However* 


USING  A  FREQUENCY  STANDAKD 


11-07 


the  accuracy  may  still  be  very  high,  because  the  beat  frequency  can  be  made  such  a  very 
small  part  of  the  total  range,  thus  reducing  the  necessity  for  great  accuracy  in  its  measure- 
ment. 

STROBOSCOPIC  METHODS.  Stroboscopic  methods  are  used  for  comparing  the 
rate  of  one  mechanical  rotation  or  vibration  with  another  or  with  the  frequency  of  a  fluc- 
tuating source  of  illumination.  They  are  essentially  low-frequency  methods  but  have 
been  used  to  measure  speeds  of  rotation  in  excess  of  10,000  rps.  The  accuracy  of  compar- 
ison is  limited  only  by  the  constancy  of  the  rates  involved  and  the  duration  of  a  measure- 
ment, being  in  that  respect  like  the  beat  methods  just  discussed. 

In  general  a  means  is  provided  for  permitting  an  observer  to  see  an  object  periodically 
for  very  short  intervals  of  time.  If  the  object  has  a  periodic  structure  like  gear  teeth  or 
spokes,  and  if  in  the  interval  between  glimpses  it  rotates  a  distance  equal  to  one  or  any 
whole  number  of  elements,  it  will  appear  to  be  stationary.  If  the  glimpse  frequency  is 
not  exactly  equal  to  the  apparent 
periodicity  in  the  rotating  struc- 
ture, it  will  appear  to  move  slowly, 
the  amount  and  direction  of  mo- 
tion corresponding  to  the  relative 
rates.  The  time  required  for  an 
apparent  motion  of  the  structure 
equal  to  one  element  space  may  be 
considered  as  the  duration  of  one 
"beat"  and  treated  as  such  in  the 
comparison  of  rates.  This  method 
can  be  used  in  a  large  number  of 
ways  for  measuring  or  comparing 
rotation  speeds  or  for  measuring 
frequency  in  terms  of  known  rota- 
tion speeds,  or  vice  versa. 

A  simple  apparatus  for  either 
case  consists  of  a  neon  or  other  va- 
por lamp  which  can  be  flashed  pe- 
riodically by  pulsating  current,  and 
a  disk  such  as  shown  in  Fig.  6  hav- 
ing concentric  rows  of  black  and 
white  sectors  of  different  periodic- 
ity, attached  to  a  suitable  rotating 
mechanism.  If  properly  chosen, 
one  row  of  sectors  will  appear  sta- 
tionary or  nearly  so  in  the  intermittent  illumination,  and  from  the  observation  the 
ratio  of  the  rotation  speed  to  the  flash  frequency  can.  be  deduced  readily  and  with  great 
accuracy. 

Similarly  the  frequency  of  any  mechanical  vibration  of  sufficient  amplitude  can  be 
determined  by  observing  the  motion  with  a  light  flashed  at  a  frequency  that  can  be  ad- 
justed to  the  proper  range.  If  the  flash  frequency  is  adjusted  to  the  highest  value  that 
will  give  a  single  image  of  the  vibrating  part,  that  is  the  desired  frequency,  which  can 
then  be  measured  by  electrical  means  or  read  from  a  calibrated  dial. 

CATHODE-RAY  OSCILLOSCOPE.  Probably  the  most  useful  of  aH  laboratory 
apparatus  for  frequency  comparison  is  the  cathode-ray  oscilloscope,  now  available  in 
many  compact  and  convenient  forms.  (See  also  Section  15,  article  23.) 

The  oscilloscope,  illustrated  in  Fig.  7,  consists  of  a  device  for  producing  a  stream  of 
electrons  which  is  directed  toward  a  fluorescent  screen  within  an  evacuated  tube,  and  of 
means  for  deflecting  the  electron  beam  in  accordance  with  currents  or  voltages  to  be 
studied,  thereby  moving  the  luminous  spot  on  the  screen.  At  ordinary  frequencies  the 
motion  of  the  spot  is  so  rapid  that  its  path  is  indicated  by  a  continuous  trace, 

In  the  usual  form  of  tube  the  deflections  are  obtained  electrostatically  by  applying 
voltage  to  pairs  of  parallel  electrodes  between  which  the  electron  beam  passes.  When  two 
voltage  waves  are  to  be  compared  they  are  connected  to  the  two  pairs  of  mutually  perpen- 
dicular plates  corresponding  to  the  x  and  y  axes  in  a  cartesian  coordinate  system,  j  The 
resulting  motion  of  the  spot  on  the  screen  is  such  that 


.  6.    Stroboscope  Disk  (Courtesy  of  General  Radio  Co.) 


x  = 


-f- 


y  =  &eo  sin  (a$  -f  <ps) 

where  k  is  a  constant  for  the  tube  and  where  e,  co,  and  <p  are  the  voltage,  angular  velocity, 
and  phase  corresponding  to  the  input  waves. 


11-08 


ELECTRICAL  MEASUREMENTS 


The  actual  figure  that  is  traced  can  be  determined  analytically  in  simple  cases  by  elim- 
inating t  between  the  two  equations.    For  example,  if  ei  =  ez,  «i  =  «2,  and  <&  —  <pi  =  ?r/2, 


Electron 
Bearr 


_ 

-/7nrzr>- 

m 


•    J 

Fio.   7.     Schematic  of  Carfiode-ray  Oscilloscope,  Illustrating  Electrostatic   Method  for  Obtaining 

lissajous  Figures 

we  get  x2  4-  y2  =  &V,  which  is  the  equation  of  a  circle.  As  the  angle  ((pz  —  <pi)  changes 
we  get  various  phases  of  an  ellipse  until,  when  it  becomes  0  or  TT,  it  degenerates  into  a 
straight  line  inclined  45°  to  the  axes.  Thus  when  two  waves  are  compared  having  equal 
amplitude  and  nearly  equal  frequencies,  the  pattern  goes  through  a  complete  series  of 
ellipses  once  for  each  cycle  gained  or  lost  by  one  frequency  on  the  other.  One  such  cycle 


Phase  Angle 


Musical 
Internal 


90° 


135° 


Dnlson 


Octave 


Octave  and  Fifth 


Frequency 

RaiTo 

1.1 


2.1 


3.1 


3.2 


Fourth 


FIG.  8.     Typical  Lissajous  Figures  with  Corresponding  Frequency  Ratios  and  Musical  Intervals 

of  pattern  changes  corresponds  to  one  "beat"  between  the  input  frequencies.    Measuring 

the  frequency  of  recurrence  of  such  beats  provides  a  convenient  and  extremely  accurate 

comparison  between  the  input  frequencies. 

Patterns  formed  in  this  way  by  frequencies  that  are  equal  or  related  as  m  I  n,  where  m 

and  n  are  integers,  are  known  as  Lissajous  figures.    Several  such  figures  corresponding  to 

simple  frequency  ratios  and  different 
phase  angles  are  shown  in  Fig.  8. 

When  frequencies  are  to  be  com- 
pared which  differ  by  a  rather  large 
ratio,  say  10  :  1,  a  figure  is  obtained 
such  as  shown  in  Fig.  9.  The  figure 
looks  like  the  projection  of  a  sinusoidal 
trace  of  10  cycles  around  a  transparent 
cylinder.  For  illustration  only,  the 
drawing  shows  half  of  the  figure  solid, 


FIG.  9.    Ten-to-one  Lissajous  Figure 


corresponding  to  the  front  side  of  the  hypothetical  cylinder,  and  the  remaining  half  dotted. 
Actually,  as  viewed  on  the  screen  these  two  parts  are  indistinguishable  in  appearance. 
If  the  ratio  departs  very  slightly  from  10  :  1,  the  pattern  moves  as  though  the  hypothetical 


USING  A  FREQUENCY  STANDARD  11-09 

cylinder  rotated  slowly  at  the  rate  of  1  complete  revolution  for  every  cycle  gained  or  lost 
at  the  low  frequency.  Thus  when  the  part  shown  solid  in  Fig.  9  moves  to  the  right, 
the  part  shown  dotted  moves  in  the  opposite  direction. 

In  order  to  avoid  the  confusion  of  this  double  pattern,  most  cathode-ray  oscilloscopes 
are-  now  equipped  with  sawtooth  wave  sweep  circuits,  the  frequency  of  which  may  be 
precisely  controlled  by  an  external  standard.  By  this  means  the  back  pattern  is  eliminated 
and  the  whole  visible  pattern  stands  or  moves  as  a  unit. 

Sometimes  it  is  desirable  to  compare  frequencies  bearing  a  large  ratio  such  as  100  '  1, 
or  even  1000  :  1,  in  order  to  study  small  departures  from  such  ratios,  determined  by  other 
means,  or  to  adjust  the  frequencies  to  those  exact  ratios.  The  patterns  corresponding  to 
such  large  ratios  are  too  complex  to  be  used  in  determining  them  directly,  but  by  means 
of  an  auxiliary  oscillator  of  intermediate  frequency  it  is  easy  to  obtain  a  nearly  exact 
adjustment  in  two  or  three  stages.  For  example,  the  100  :  1  setting  can  be  obtained  by 
means  of  an  auxiliary  oscillator  having  10  times  the  lower  of  the  two  frequencies  concerned. 
As  soon  as  the  approximate  adjustment  is  obtained  between  the  end  frequencies,  the  inter- 
mediate oscillator  can  be  dispensed  with  and  the  final  comparison  carried  out  with  the 
large  ratio  pattern.  In  such  a  procedure  the  horizontal  gain  of  the  oscilloscope  may  be 
made  so  large  that  only  part  of  the  complete  figure  appears  on  the  screen.  Because  of  the 
enlargement  of  the  figure  obtained  in  this  way  the  accuracy  of  observation  is  materially 
improved.  This  is  essentially  a  zero-beat  method  permitting  extreme  accuracy  of  com- 
parison. 

Frequencies  over  a  very  wide  range  may  be  compared  by  the  cathode-ray-oscilloscope 
method,  since  many  oscilloscope  tubes  will  produce  clear  figures  with  inputs  from  the 
lowest  frequencies  up  to  many  megacycles.  The  interpretation  of  multiline  figures  corre- 
sponding to  fractional  ratios  and  many  special  methods  for  using  the  oscilloscope  are 
described  in  the  references. 

AUDIBLE  METHODS.  Audible  methods  are  often  useful,  especially  as  a  means  for 
observing  in  one  of  the  various  beat  methods.  For  example,  if  two  frequencies  which  are 
nearly  alike  can  be  heard  simultaneously,  the  beat  frequency  may  be  sensed  as  variations 
in  loudness.  The  beats  may  be  counted  in  order  to  determine  the  departure  of  one  fre- 
quency from  the  other,  or  one  of  the  frequencies  can  be  adjusted  to  match  the  other  by 
listening  for  zero  beat.  If  one  or  both  of  the  sound  sources  is  rich  in  harmonics  other  ratios 
than  1  i  1  may  be  studied  readily  by  this  means. 

When  two  audible  frequencies  are  nearly  alike  and  one  source  is  movable  in  actual 
location,  it  is  sometimes  convenient  to  use  the  Doppler  effect  to  determine  which  one  is 
high.  For  example,  if  an  observer  holds  a  vibrating  tuning  fork  in  a  stationary  sound 
field  of  nearly  the  same  frequency,  slow  beats  will  indicate  the  magnitude  but  not  the  sign 
of  the  frequency  difference.  However,  if,  while  still  vibrating,  the  fork  is  moved  away 
from  the  observer  and  the  beat  frequency  becomes  lower  (while  moving) ,  the  fork  fre- 
quency is  higher  than  that  of  the  sound  field. 

Often  the  musical  pitch  sense  may  be  used  to  advantage  in  comparing  the  actual  pitch 
of  two  tones  one  of  which  may  be  considered  as  standard.  Since  a  musical  semitone  is 
approximately  6  per  cent,  it  is  evident  that  by  this  means  alone  the  ratio  of  two  tones  in 
the  musical  range  may  be  estimated  to  well  within  5  per  cent.  As  a  direct  measurement, 
this  accuracy  is  insufficient  for  most  purposes,  but  it  is  good  enough  to  be  very  useful  in 
making  estimates  of  beat  frequencies  between  two  high-frequency  sources.  For  example, 
a  5  per  cent  error  in  the  500-cycle  beat  between  two  frequencies,  nominally  25  megacycles, 
corresponds  to  an  error  of  only  1  part  in  a  million  in  their  comparison. 

As  an  aid  to  this  method  it  is  convenient  to  keep  in  mind  that  the  frequency  ratios 
corresponding  to  the  consecutive  pairs  of  notes  in  the  major  diatonic  scale  are: 

9         10        16        9         10        9         16 
8         9         15        8         9         8         15 

The  product  of  all  these  together  equals  2,  that  is,  an  octaxe.  Also  the  product  of  the 
first  four  equals  3/a,  that  is,  the  musical  fifth,  and  similarly  for  the  other  recognized  musical 
intervals. 

Most  keyboard  musical  instruments  are  tuned  to  the  equally  tempered  scale  in  which  the 

interval  corresponding  to  all  semitones  is  equal  to  V  2,  and  a  whole  tone  equals  two  semi- 
tones. In  Table  2  the  actual  frequencies  are  listed  for  a  range  including  that  of  an  88  note 
piano  with  A  above  middle  C  equal  to  440  vibrations  per  second.  This  is  the  most  generally 
accepted  standard  of  musical  pitch.  For  the  convenience  of  using  round  numbers,  a  pitch 
system  is  sometimes  specified  in  which  all  the  C's  are  powers  of  2,  middle  C  being  256 
vibrations  per  second.  The  frequencies  in  this  scale  are  about  2  per  cent  lower  than  in 
the  accepted  standard  of  musical  pitch,  corresponding  to  about  a  third  of  a  semitone.  The 


11-10 


ELECTRICAL  MEASUREMENTS 


Table  2.    Equally  Tempered  Scale  A  =  440 


C4-C3 

Cr-C2 

CrOi 

Ci-C 

C-C' 

C'-C2 

C2-C3 

C3-04 

c 

16.35 

32.70 

65.41 

130.81 

261.63 

523.25 

1046.5 

2093.0 

c# 

17.32 

34.65 

69.30 

138.59 

277.18 

554.37 

1108.7 

2217,5 

D 

18.35 

36.71 

73.42 

146.83 

293.66 

587.33 

1174.7 

2349.3 

D* 

19.45 

38.89 

77.78 

155.56 

311.13 

622.25 

1244.5 

2489.0 

B 

20.60 

41.20 

82.41 

164.81 

329.63 

659.25 

1318.5 

2637.0 

F 

21.83 

43.65 

87.31 

174.61 

349.23 

698.46 

1396.9 

2793.8 

F# 

23.12 

46.25 

92.50 

185.00 

369.99 

739.99 

1480.0 

2960.0 

G 

24.50 

49.00 

98.00 

196.00 

392.00 

783.99 

1568.0 

3136.0 

G# 

25.96 

51.91 

103.83 

207.65 

415.30 

830.61 

1661.2 

3322.4 

A 

27.50 

55.00 

110.00 

220.00 

440.00 

880.00 

1760.0 

3520.0 

A£ 

29.14 

58.27 

116.54 

233.08 

466.16 

932.33 

1864.7 

3729.3 

B 

30.87 

61.74 

123.47 

246.94 

493.88 

987.77 

1975.6 

3951.1 

C 

32.70 

65.41 

130.81 

261.63 

523.25 

1046.50 

2093.0 

4186.0 

PIG.      10.     Frequency      Comparison 
with  Interpolating  Oscillator 


latter  pitch  is  used  chiefly  in  physics  and  sometimes  is  known  as  physical  pitch.     Inter- 
national pitch  is  based  on  A  =  435. 

INTERPOLATION  METHODS.  Most  measurements  of  frequency,  apart  from  the 
inter  comparison  of  standards  and  similar  studies,  can  be  made  most  expediently,  and  with 
sufficient  accuracy,  by  the  use  of  a  calibrated  interpolating  oscillator  in  combination  with 
some  means  such  as  just  described  for  indicating  exact  frequency  relationships.  The 
principle  is  illustrated  in  Fig.  10,  which  may  be  modified  or  extended  in  numerous  ways 
depending  on  the  application.  It  is  evident  that  other  types  of  indicator  than  oscilloscopes 

could  be  used,  and  that  by  means  of  suitable  switches, 
JSi  and  $2,  only  one  indicator  is  needed  for  the  simple 
example  described  in  the  following. 

The  operation  is  best  explained  by  an  example.  Let 
the  standard  be  100,000  cycles,  and  let  us  determine  a 
frequency,  say  3,564,000  cycles,  by  means  of  an  inter- 
polating oscillator  stable  in  the  range  from  500,000  to 
600,000  cycles  and  having  a  vernier  dial,  FL,  calibrated 
in  this  range.  By  a  little  experimenting  one  finds  readily 
that  an  integral  6  :  1  pattern  is  observed  on  oscilloscope 
B  when  the  interpolating  oscillator  is  set  a  little  below 
600,000.  That  is  the  cue  to  calibrate  the  interpolating 
oscillator  at  600,000  in  terms  of  the  standard  by  setting 
the  dial  to  that  exact  value  and  adjusting  vernier  V$ 
(which  adjusts  the  frequency  slightly,  independently  of 
FI)  until  the  ratio  of  the  pattern  on  oscilloscope  A 
is  exactly  6:1.  When  the  oscillator  is  again  adjusted  to  give  a  6  1  1  stationary  pattern 
on  oscilloscope  B,  the  dial  reading  multiplied  by  6  will  be  the  frequency  to  be  determined. 
In  general  it  is  best  to  so  calibrate  the  interpolating  oscillator  that  the  two  positions  on 
the  calibration  dial  are  as  near  together  as  possible.  When  the  oscilloscope  is  the  indicator 
it  is  often  desirable  to  use  multiline  figures  in  making  one  or  both  settings  in  order  to  accom- 
plish this.  For  example,  if  the  unknown  frequency  bore  some  simple  ratio  to  533,000 
cycles,  the  three  line  pattern  corresponding  to  the  ratio  5  lfo  should  be  used  in  calibrating 
the  interpolating  oscillator.  It  is  evident  that  the  overall  accuracy  of  the  general  method 
may  be  increased  materially  by  this  means  especially  if  an  auxiliary  vernier  dial  is  provided 
which  is  calibrated  in  a  small  percentage  range  and  which  can  be  used  to  interpolate  be- 
tween two  close  settings. 

The  method  illustrated  in  Fig.  10  may  be  used  in  setting  the  "unknown"  frequency  at 
any  value  P  X  Q  times  the  standard,  where  P  and  Q  are  integers  or  simple  rational  frac- 
tions. Using  stationary  figures  on  both  oscilloscopes  simultaneously,  the  accuracy  of 
setting  is  very  high.  The  * 'unknown"  frequency  can  then  be  used  as  standard  in  a  wide 
choice  of  frequencies  to  extend  the  range.  Sometimes  it  is  convenient  to  use  more  than 
one  interpolating  oscillator  when  it  is  necessary  to  cover  a  wide  range  of  frequencies. 

In  particular  this  method  can  be  used  for  extending  the  frequency  of  a  standard  upward 
for  the  purpose  of  measuring  at  ultra  high  frequencies  in  the  decimeter  and  centimeter 
range.  As  yet,  however,  the  oscilloscope  cannot  be  used  to  obtain  observable  Lissajous 
figures  at  the  highest  frequencies  because  the  stability  of  such  oscillators  is  not  yet  good 
enough  to  produce  stationary  figures.  This  may  be  expected  to  come  with  time,  however, 
and  already  cathode-ray  tubes  have  been  produced  capable  of  resolving  single  traces  at 
frecjueaeies  as  high  as  10,000  megacycles. 


EMPLOYING  CIRCUIT  ELEMENT  SELECTIVITY        11-11 


For  ultra-high-frequency  measurements  in  terms  of  a  standard  the  usual  method  is  to 
heterodyne  a  harmonic  of  a  measurable  source,  such  as  the  "unknown"  of  Fig.  10,  -with 
the  high  frequency  by  means  of  a  crystal  detector  and  either  to  estimate  the  frequency 
of  the  relatively  low-frequency  components  obtained  or  to  change  the  variable  source  until 
the  detector  output  is  as  near  to  zero  frequency  as  can  be  estimated.  This  does  not  have 
to  be  actually  zero  to  be  good;  it  should  be  remembered  that,  when  measuring  10,000 
megacycles,  corresponding  roughly  to  3-cm  waves,  10,000  cycles  in  the  beat  corresponds  to 
only  1  part  in  a  million  in  the  overall  measurement. 

Standards  of  frequency  of  very  great  accuracy  are  made  available  by  the  National 
Bureau  of  Standards  through  continuous  radio  transmissions  from  station  W  WV.  All 
the  carrier  frequencies,  which  are  2.5,  5,  10,  15,  20,  25,  30,  and  35  megacycles,  are  regu- 
lated by  the  primary  frequency  standard  of  the  bureau.  Each  carrier  is  modulated  with 
seconds  pulses  and  with  the  audio  frequencies  440  and  4000  cycles,  also  of  very  high 
accuracy. 

4.  EMPLOYING  CIRCUIT  ELEMENT  SELECTIVITY 

When  extreme  accuracy  is  not  a  primary  requirement,  or  when  standard-frequency 
current  is  not  available,  it  is  often  convenient  to  measure  frequency  in  terms  of  the  response 
of  selective  electrical  networks  or  mechanical  resonators. 

METERS  FOR  POWER  FREQUENCIES.  Most  of  the  commercial  meters  for  indi- 
cating power  frequency  operate  on  one  of  three  principles.  Meters  of  the  reed  type  employ 
a  number  of  reeds  tuned  consecutively  to  slightly  different  values  and  loosely  coupled  to 
an  electromagnet  energised  by  the  current  to  be  measured.  The  reeds  whose  frequencies 


-i  - 

v 

J->        ^v       ^ 

J  J 

t        » 

\ 

1 

1 

A.C. 

\ 

1    Input 

I 
I 

V 

One  Reed""i 

in  Vibration         ' 

'&AVW 

58 


59 


60 

1 


61 

1 


62 

1 


Appearance  of  Scale 

When  Indicating 

Frequency  of  Input 


FIG.  11.     Vibrating-reed  Frequency  Meter 

correspond  most  nearly  to  the  frequency  of  the  input  current  vibrate  at  the  largest  ampli- 
tude. Usually  the  reeds  are  arranged  in  a  row,  as  shown  in  Fig.  11,  with  the  ends  in  line 
near  a  scale  so  that  the  frequency  of  the  reed  with  greatest  amplitude  can  be  read  oft 
directly.  Such  meters  are  available  in  a  considerable  range  of  frequencies  and  are  very 
useful  for  measuring  low  frequencies  directly  or  for  indicating  beat  frequencies  in  their 
range. 

The  Weston  frequency  meter,  shown  in  Fig.  12,  has  a  movable  soft-iron  armature  free 
to  move  in  the  resultant  field  from  two  mutually  per- 
pendicular coils.  When  the  frequency  has  some  nomi- 
nal value,  the  fields  in  the  two  coils  are  equal  and  the 
armature  takes  up  a  nominal  position  parallel  to  the 
resultant  field.  When  the  frequency  changes,  the 

ratio  of  currents  in  the  coils  changes, 

owing  to  the  frequency  selectivity  of 

the  input  circuits,  causing  a  shift  in 

the  resultant  field  and  a  correspond- 

ing movement  of  the  armature. 
The  induction  frequency  meter, 

shown  in  Fig.  13,  consists  essentially 

of  two  opposing  induction  voltmeter 

elements  associated  with  one  arma- 

ture.    The  two  motor  elements  are 

suppled  through  resistive  and  reac- 

tive  circuits  respectively  so  that  the 

ratio  of  the  effective  currents  varies 
with  the  applied  frequency.    In  the  type  of  meter  shown,  a  circular  disk  is  used  and  the 
indicating  position,  is  that  for  which  a  restoring  spring  just  balances  the  resultant  torque 


FIG.  12.    Weston 
Frequency  Meter 


13     induction  Frequency  Meter 


11-12 


ELECTEICAL  MEASUREMENTS 


from  the  opposing  drive  elements.  The  deflection  is  therefore  proportional  to  the  frequency 
deviation  from  a  nominal  value. 

In  another  type  of  this  meter  no  restoring  spring  is  used,  but  the  armature  is  so  shaped 
for  one  or  both  motor  elements  that  the  torque  varies  with  angle  of  deflection  as  well  as 
with  input  current.  This  permits  an  angular  balance  position  to  be  obtained  which  does 
not  depend  upon  the  applied  voltage  or  the  reaction  of  a  spring. 

Some  meters  for  power  frequencies  employ  resonance  to  increase  the  sensitivity  in  a 
narrow  frequency  range.  These  and  others  are  described  in  standard  works  on  power 
meters. 

BRIDGE  METHODS.  Various  bridge  methods  may  be  used  for  measuring  frequency. 
The  one  shown  in  Fig.  14  is  typical.  The  bridge  is  balanced  when, 


With  head  phones,  or  other  suitable  null  indicator,  the  frequency  can  be  measured  in 

terms  of  Rit  R%,  Ci,  and  CV    As  a  frequency  meter  two  of  these  elements  can  be  made 

variable  and  calibrated  to  read  input  frequency  at 
balance.  For  more  detail  about  bridge  measuring 
devices  see  below,  article  12. 

THE  MONO  CHORD.  This  is  a  useful  labora- 
tory tool  for  the  approximate  determination  of  fre- 
quency in  the  lower  and  medium  audible  ranges. 
It  consists  of  a  steel  wire  under  tension  between 
movable  bridges,  the  wire  being  coupled  electro- 
magnetically  to  the  a-c  input  to  be  measured.  The 
frequency  of  maximum  response  can  be  determined 
from  the  constants  of  the  system,  or  a  calibration 
can  be  made  in  terms  of  length  or  tension.  Under 

FIG.  14.    Bridge  for  Measuring  Frequency   some  conditions  an  accuracy  of  1  part  in  1000  may 

be  obtained  with  this  means. 
RADIO  WAVEMETER.    The  tuned  circuit  wavemeter,  Fig.  15,  used  extensively  for 

radio-frequency  measurements  from  10  kc  to  100,000  kc,  consists  primarily  of  a  coil  and 

condenser  connected  in  a  closed  circuit  and  loosely  coupled  to~  a  source  to  be  measured. 

The  circuit  is  tuned,  usually  by  means  of  the  condenser,  until  a  resonance  condition  is 

indicated.    From  a  scale  on  the  condenser,  previously  calibrated  by  means  of  accurately 

known  input  frequencies,  the  frequency  of  any  source  in  a  limited  range  can  be  read  off 

directly.    In  commercial  wavemeters  of  this  sort  a  set  of  coils  is  generally  provided  suitable 

for  use  in  a  number  of  adjacent  and  somewhat  overlapping  ranges. 

The  coupling  from  the  source  to  be  measured  may  be  effected  (1)  through  a  low  im- 

pedance, such  as  a  low  resistance,  in  series  with  the  tuned  circuit;  (2)  by  loose  magnetic 

coupling;  or  (3)  by  loose  capacitance  coupling  as  indicated  in  Fig.  15. 

Resonance  may  be  indicated  (1)  by  a  current-indicating  instrument,  such  as  a  thermal 

galvanometer,  in  series  with  the  timed  circuit;  (2)  by  a  voltage-indicating  instrument,  such 

as  a  crystal  detector  or  diode  in 

parallel  with  the  reactive  elements 

(if  the  source  is  modulated  by  au- 

dio frequency,  head  phones  may 

provide  the  most  convenient  means 

for  observing)  ;  (3)  by  the  reaction 

on  a  power-indicating  means  asso- 

ciated with  the  source;  (4)  by  a 

measure   of  power  in  a  separate 

aperiodic   circuit    coupled   to   the 


7 


Shield' 


FIG.  15.     Radio  Wave  Meter.     Resonance  type  with  high 
impedance  resonance  indicator. 


tuned  circuit  but  not  directly  to 
the  source;  and  (5)  by  means  of 
amplification  followed  by  detection 
as  in  a  simple  radio  receiver. 

Although  all  five  methods  are  used,  the  fifth  is  preferable  because  of  the  vanishingly 
small  effect  of  the  indicating  means  on  the  Q  of  the  tuned  circuit  and  hence  on  the  precision 
of  observation.  Used  as  indicated,  an  effective  Q  of  500  may  be  attained  in  a  good  part 
of  the  range,  and  with  good  equipment  and  careful  procedure  an  accuracy  of  the  order 
of  1  part  in  104  may  be  achieved. 

QUARTZ  RESONATORS.  Specific  frequencies  may  be  indicated  with  great  accuracy 
by  means  of  quartz  resonators  coupled  loosely  across  the  tuning  elements  of  a  resonant 
wavemeter.  Owing  to  the  relatively  very  much  higher  Q  of  the  crystal,  which  is  often  in 


EMPLOYING  CIRCUIT  ELEMENT  SELECTIVITY        11-13 

excess  of  100,000,  its  response  characteristic  is  confined  to  a  correspondingly  narrow  and 
well-defined  range  of  frequency.  This  response  characteristic  is  superposed  on  the  broader 
characteristic  of  the  tuned  circuit.  Used  in  this  way  such  resonators  are  valuable  chiefly 
in  fixing  calibration  points  on  the  wavemeter.  A  variable  input  can  be  adjusted  until 
the  crystal  response  is  indicated;  then,  with  the  same  input  frequency,  the  wavemeter 
condenser  can  be  adjusted  for  maximum  response  to  fix  the  calibration  at  the.frequency 
of  the  crystal. 

Quartz  crystals  may  be  used  in  a  great  variety  of  ways  as  single-frequency  indicators. 
One  interesting  adaption,  due  to  Giebe  and  Scheibe,  consists  of  a  series  of  resonators 
mounted  in  a  gas  at  low  pressure  which  may  be  coupled  electrostatically  to  a  circuit  to 
be  tested.  If  a  frequency  is  applied  which  corresponds  to  that  of  one  of  the  resonators, 
the  large  potential  gradients  in  the  neighborhood  of  that  resonator  due  to  its  resonance 
will  cause  a  luminous  discharge  which  serves  as  indicator.  This  method  is  somewhat 
analogous  to  the  reed  indicator  but  is  applicable  to  frequencies  up  to  a  million  cycles  per 
second  or  more  and  of  course  is  inherently  much  more  accurate. 

CAVITY  RESONATORS.  The  most  convenient  and  generally  satisfactory  means  for 
measuring  frequencies  in  the  region  from  200  to  30,000  megacycles  has  been  cavity 
resonators,  which  have  been  developed  in  a  variety  of  forms  for  different  frequency  ranges 
and  for  different  methods  of  use.  A  cavity  is  primarily  an  almost  completely  enclosed 
space  in  a  rigid  piece  of  metal,  with  openings  only  for  coupling  electromagnetic  energy 
from  wave  guides  or  coaxial  conductors,  and  usually  also  for  the  operation  of  a  plunger 
for  frequency  adjustment. 

The  cavity  itself  in  its  lowest-frequency  modes  may  be  considered  a  familiar  tuned 
circuit  in  which  the  capacitance  is  formed  by  opposite  sides,  sometimes  the  two  flat  ends 
of  a  cylinder,  and  connected  by  a  continuous  array  of  single-turn  coils  forming  the  cylinder 
wall.  It  is  evident  that  for  a  moderate-sized  cavity  the  effective  capacitance  and  induct- 
ance are  both  very  small  and  hence  the  frequency  is  very  high.  Since  there  are  no  radiation 
losses,  and  since  the  interior  losses  may  be  kept  small,  the  Q  factor  may  be  very  high.  For 
silver-plated  fixed  cavities  the  Q  may  exceed  20,000;  for  adjustable  cavities  it  is  somewhat 
less. 

Cavities  are  resonant  elements  and  as  such  may  be  used  as  selective  transmission  devices 
indicating  a  maximum  of  transmitted  energy  into  a  detector,  or  as  selective  absorption 
devices  indicating  a  minimum  in  a  high-impedance  source.  In  either  case  the  actual 
detection  is  most  easily  accomplished  by  a  crystal  detector  which  may  actuate  a  d-c  meter, 
or,  if  the  source  is  modulated,  a  set  of  head  phones  may  be  used,  with  amplification  if 
necessary.  The  setting  accuracy  may  be  as  high  as  1  part  in  105. 

The  design,  construction,  and  use  of  cavity  resonators  are  discussed  at  length  in  Section 
7  and  in  references. 

TRANSMISSION  LINES.  Very  high  frequencies  can  be  measured  with  fair  accuracy 
by  a  study  of  standing  waves  in  transmission  lines,  the  three  usual  types  being  Lecher  wires, 
coaxial  conductors,  and  wave  guides.  The  simplest  method  involves  a  movable  short- 
circuiting  conductor  which  can  be  moved  along  the  line  by  external  control,  and  the  ob- 
servation of  successive  minima  due  to  reaction  on  the  source  of  high-frequency  energy  fed 
into  the  line.  The  actual  distance  between  successive  positions  of  the  short-circuiting 
conductor  which  causes  such  minima  determines  a  half  wavelength  for  the  particular  line 
used.  In  the  case  of  Lecher  wires  (parallel  wires  a  few  centimeters  apart)  and  coaxial 
lines,  the  frequency  is  given  approximately  by 

3  X  1010 
•  2Z 

where  I  equals  the  distance  between  resonance  positions.  A  calibrated  instrument  based 
on  this  method  may  be  accurate  to  about  0.1  per  cent. 

Although  Lecher  wires  are  convenient  for  purposes  of  demonstration,  a  low-resistance 
lamp  on  the  short-circuiting  rider  being  a  suitable  indicator  of  resonance,  the  attainable 
accuracy  is  limited  by  large  radiation  losses,  especially  at  high  frequencies.  Wave  guides 
are  inconvenient  for  this  purpose  on  account  of  the  wide  variety  of  modes  that  may  give 
false  indications  unless  special  precautions  are  taken  to  suppress  them.  High-frequency 
transmission  lines  are  discussed  in  Section  10,  and  some  of  the  accompanying  references 
deal  with  their  use  as  frequency-measuring  devices. 

THE  ECHELETTE  GRATING.  A  method,  closely  analogous  to  the  familiar  optical 
grating,  has  been  developed  for  establishing  the  frequency  of  the  shortest  electromagnetic 
radiations  producable  experimentally.  The  method  consists  in  directing  the  radiation  in 
question  toward  an  array  of  stepped  metal  reflectors,  like  the  slats  in  a  Venetian  blind,  and 
in  observing  the  angle  of  reflection  for  which  the  greatest  intensity  of  reflected  energy  is 
obtained.  From  the  spacing  of  the  reflectors,  and  the  measured  angles  for  maximum 


11-14 


ELECTBICAL  MEASUREMENTS 


energy,  the  wavelength  may  be  computed  with  considerable  accuracy  as  with  the  optical 
grating.  This  method  of  measurement,  in  fact,  was  an  important  step  in  establishing  the 
continuity  of  the  electromagnetic  spectrum  between  light  produced  by  thermal  excitation 
and  electric  waves  produced  by  purely  electrical  means. 


5.  ELECTROMAGNETIC  PHENOMENA 

Table  3  shows  in  a  graphical  manner  the  relation  between  the  frequencies  of  various 
observed  electromagnetic  phenomena.  Where  any  considerable  uncertainty  exists  as  to 
the  extent  of  a  range  due  to  ambiguity  of  definition  or  due  to  disagreement  among  sources, 
the  uncertainty  is  indicated  by  dotted  extensions. 

Table  3.    Frequency  Spectrum  of  Electrical  Phenomena 


Cosmic  rays 


Q&Ql  A-s- 


1  x-imli 


1  angstrom  ttnit 


Sodium  D  Fines 


Highest  frequency  produced  electrically 


Highest  frequency  continuous  wave 


National  Bureau  of  Standards 
Standard  frequency  transmissions — j 
2.5, 5,  10, 15, 20,  25, 30,  and  35  Me  L 


MiddJeC 


Seconds 


radl-afiOQ 
produced  electrically 


ommercial  radio 
ndard  broadcasting 


--  Audible  range 

l  telephone 


j-  Audible  rang 
rComrnercial 


1QJ^_         | 

» 1  n5 —        ! 


D-c  telegraph 


The  lower  end  of  the  chart,  and  extending  a  little  beyond  3  X  1010,  includes  the  entire 
range  over  which  continuous  alternating  current  can  be  produced  at  the  present  tune. 
This  is  also  the  entire  range  over  which  frequencies  can  be  measured  or  compared  directly. 
Damped  waves  having  frequencies  up  to  3  X  1012  have  been  produced  by  purely  electrical 
methods.  In  this  region,  which  overlaps  the  infrared  spectrum,  the  frequencies  can  be 
determined  by  either  electrical  resonance  methods  or  by  optical  interference  methods. 

From  1Q12  upwards  the  terms  on  the  right  refer  to  electromagnetic  waves  originating  in 
hot  or  otherwise  luminous  bodies,  in  excited  atoms,  or  in  atom  nuclei.  Between  1012 
and  1Q20  the  frequencies  are  calculated  from  measured  wavelengths;  from  10s*  upward 
they  are  calculated  from  measured  photon  energies.  In  certain  ranges  either  of  two  names 
is  applicable,  as  indicated  by  the  overlapping  grouping. 

It  is  interesting  to  note  that,  by  extendiing  the  frequency  spectrum  downward,  frequen- 
cies of  recurring  astronomic  phenomena  are  encountered  that  are  about  as  far  removed 
from  onje  cycle  as  are  the  highest  frequencies  noted.  Thus  the  frequency  of  rotation  of 
tibe  earth  on  its  axis  is  1.16  X  10~5;  the  frequency  of  revolution  of  the  earth  around  the 
SOB  is  3.17  X  10"8.  The  frequency  of  rotation  of  the  equinoxes  around  the  ecliptic  is 


BIBLIOGRAPHY  11-15 

1.2  X  10""12.  The  frequency  of  rotation  of  Andromeda  Nebula  is  estimated  (Jeans)  to 
be  1.7  X  10~15,  and  it  is  perhaps  reasonable  to  suppose  that  frequencies  of  periodic  phe- 
nomena involving  interactions  between  the  nebulae  may  be  many  orders  smaller.  Thus 
the  frequencies  to  which  the  senses  respond  most  readily  are,  on  a  logarithmic  scale,  about 
midway  between  the  extremes  of  which  we  are  aware. 

BIBLIOGRAPHY 

Books 

Jansky,  Cyril  M.,  Electrical  Meters,  McGraw-Hill,  1917. 

Pierce,  George  W.,  Electric  Oscillations  arid  Electric  Waves,  McGraw-Hill,  1920. 
Brown,  Hugh  A.,  Radio  Frequency  Electrical  Measurements,  McGraw-Hill,  1931. 
Hund,  August,  High  Frequency  Measurements,  McGraw-Hill,  1933. 

Brainerd,  J.  G.,  Kochler,  Glen,  Reich,  Herbert  J.,  and  Woodruff,  L.  F.,  Uttra  High  Frequency  Tech- 
niques, D.  Van  Nostrand,  1942. 
Montgomery,  C.  G.,  Technique  of  Microwave  Measurements  (Rad,  Lab.  Series  11),  McGraw-Hill,  1947. 

Articles 

Cady,  W.  G.,  The  Piezoelectric  Resonator,  Proc.  I.R.E.,  Vol.  10,  No.  2,  83-114  (April,  1922). 
Nicols,  E.  F.,  and  Tear,  J.  D.,  Short  Electric  Waves,  Phys.  Rev.,  Vol.  21,  Series  II,  587-610  (June  1923). 
Pierce,  George  W.,  Piezoelectric  Crystal  Resonators  and  Crystal  Oscillators  Applied  to  the  Precision 

Calibration  of  Wave  Meters,  Proc.  Am.  Acad.  Arts  and  Sci.,  Vol.  59,  No.  4,  81-106  (October  1923). 
Hund,  August,  Theory  of  Determination  of  Ultra-radio  Frequencies  by  Standing  Waves  on  Wires, 

Scientific  Papers  of  Bureau  of  Standards  No.  491  (June  23,  1924). 
Giebe,   E.,   and   Scheibe,   A.,   Leuchtende   piezoelektrisehe  Resonatoren  als  Hochfrequenznormale, 

E.T.Z.,  Vol.  47,  380-385  (April  1926). 
Oapp,  J.  K.,  Universal  Frequency  Standardization  from  a  Single  Frequency  Standard,  J.Q.S.A.  and 

RJSJ.,  Vol.  15,  No.  1,  25-47  (July  1927). 
Rasmussen,  Frederick  J.,  Frequency  Measurements  with  the  Cathode-ray  Oscillograph,  J".  AJ.E.E., 

Vol.  46,  3-12  (July  1927). 
Allen,  George  E.,  The  Accuracy  of  the  Monochord  as  a  Measurer  of  Frequency,  Phil.  Mag,,  Vol.  4, 

Series  7,  1324-1337  (December  1927). 
Ferguson,  J.  G.,  and  Bartlett,  B.  W.,  The  Measurement  of  Capacitance  in  Terms  of  Resistance  and 

Frequency,  Bell  Sys.  Tech.  J.,  VoL  7,  42(M37  (July  1928). 

Pierce,  George  W.,  Magnetostriction  Oscillators,  Proc.  I.R.B.,  Vol.  17,  42-88  (January  1929). 
Marrison,  W.  A.,  High  Precision  Standard  of  Frequency,  BeU  Sys.  Tech.  J.,  July  1929,  pp.  493-514, 

Proc.  I.R.E.,  July  1929,  pp.  1103-1122. 
Marrison,  W.  A.,  A  Method  for  Estimating  Audible  Frequencies,  Bell.  Lab.  Rec.,  Vol.  8,  No.  4,  178- 

182  (December  1929). 
Polkinghorn,  F.  A.,  and  Roetken,  A.  A.,  A  Device  for  the  Precise  Measurement  of  High  Frequences, 

Proc.  I.R.E.,  VoL  19,  937-948  (June  1931). 
Johnson,  J.  B.r  The  Cathode-ray  Oscillograph,  J.  Franklin  Inst.,  Vol.  212,  No.  6,  687-717  (December 

1931). 
Loomis,  Alfred  L.,  and  Marrison,  W.  A.,  Modern  Developments  in  Precision  Clocks,  Trans.  AJ[JBJS.f 

June  1932,  pp.  527-537. 

Peterson,  H.  O.,  and  Braaten,  A.  M.,  The  Precision  Frequency  Measuring  System  of  R.C.A.  Communi- 
cations, Inc.,  Proc.  I.R.E.,  Vol.  20,  No.  6,  941-956  (June  1932). 
Mickey,  L.,  and  Martin,  A.  D.,  Development  of  Standard  Frequency  Transmitting  Sets,  Bur.  Standards 

J.  Research,  Research  Paper  630  (January  1934). 
Dye,  D.  W.,  and  Essen,  L.,  Valve  Maintained  Tuning  Fork  as  Primary  Standard  of  Frequency,  Proc. 

Royal  Soc.,  January  1934,  pp.  285-306. 
Forbes,  H.  C.,  and  Zaugbaum,  F.,  Frequency,  Time  Control  with  Telephone  Aid,  Elec.  World,  Jan.  20, 

1934,  p.  117. 
Seheibe,  A.,  und  Adelsberger,  U.,  Die  technischen  Einrichtungen  der  Quarzuhren  der  Physikalisch- 

Technisehen  Reichsanstalt,   Eoch  frequenztechnik   und  Elektroakustik,   February   1934,   Band  43, 

Heft  2,  pp.  37-47. 
Hazen,  Grace,  and  Kenyon,  Frieda,  Primary  Radio-frequency  Standardization  by  the  Use  of  the 

Cathode-ray  Oscillograph,  Scientific  Papers  of  Bureau  of  Standards,  No.  489  (May  1934). 
Williams,  Emrys,  Audio-frequency  Measurement  by  the  Electrically  Excited  Monochord,  Proc.  LR.E., 

VoL  22,  No.  6,  794-804  (June  1934). 
Cleeton,  Claud  E-,  Grating  Theory  and  Study  of  the  Magnetostatic  Oscillator  Frequency,  Physics, 

June  1935,  pp.  207-209. 
Cleeton,  C.  E.,  and  Williams,  N.  H.,  The  Shortest  Continuous  Radio  Waves,  Phys.  Rev.,  VoL  50, 

1091  (December  1936). 
Essen,  L.,  and  Gordon-Smith,  A.  C.,  The  Measurement  of  Frequencies  in  the  Range  100  mc/s  to  10,000 

mc/s,  J.  I.E.E.,  Part  III,  Vol.  92,  No.  20,  291-295  (December  1945).  _ 

Jones,  William  J.,  Types  and  Applications  of  Microwave  Frequency  Meters,  Radio,  January  1946, 

pp.  29-34. 
Clayton,  R.  J.,  Houldin,  J.  E.,  Lamont,  H.  R.  L.,  and  Wilkshaw,  W.  E.,  Radio  Measurements  in  the 

Decimeter  and  Centimeter  Wave  Bands,  J.  I.EJB.t  Part  III,  Vol.  93,  No.  22,  97-117  (March  1946). 
Lee,  Gordon  M.,  A  Three4>eam  Oscillograph  for  Recording  at  Frequencies  up  to  10,000  Megacycles, 

Proc.  I.R.E.,  and  Waves  and  Electrons,  VoL  34,  No,  3,  121W-129W  (March  1946). 
Essen,  L.,  Cavity  Resonator  Wave  Meters,  Wireless  Engineer ,  VoL  23,  No.  272,  126-132  (May  1946). 
Booth,  C.  F.,  and  Laver,  F.  J.  M.,  A  Standard  of  Frequency  and  Its  Applications,  J.  I.E.K,  Part  III, 

Vol.  93,  No.  24,  223-236  (Jury  1946). 

GafFney,  F.  J.,  Microwave  Measurements  and  Test  Equipment,  Proc.  I.R.S.,  and  Waves  and  Elec- 
trons, VoL  34,  No.  10,  775-793  (October  1946). 
Lafferty,  J.  M.t  A  Millimeter-wave  Ueflex  Oscillator,  J.  Applied  PAy*.,  VoL  17,  No.  12,  1061-1066 

(December  1946), 
Marrison,  W.  A.,  Tke  Evolution  of  tbe  Quartz  Crystal  Clock,  B&.TJ.,  VoL  27,  510-588  (July  194S). 


11-16 


ELECTRICAL  MEASUREMENTS 


MEASUREMENT  OF   PRIMARY  ELECTRICAL   QUANTI- 
TIES (CURRENT,  VOLTAGE,  RESISTANCE, 
CAPACITANCE,  AND  INDUCTANCE) 

By  J.  G.  Ferguson 

The  primary  quantities  of  interest  in  the  communications  field  may  be  divided  into  two 
classes:  first,  current  and  voltage;  and  second,  the  so-called  circuit  constants,  resistance, 
capacitance,  and  inductance.  Power  is  usually  obtained  from  the  measurement  of  voltage 
or  current,  and  resistance.  Frequency  is  covered  under  Frequency  Measurements,  pp. 
11-1  to  11-15. 

Values  of  current  and  voltage  generally  do  not  require  very  accurate  measurement. 
On  the  other  hand,  very  severe  requirements  are  justified  economically  in  the  design  and 
measurement  of  circuits  and  their  component  parts  if  they  result  in  an  increase  in  the 
number  of  channels  which  can  be  made  available  from  a  given  physical  circuit.  These 
requirements  are  specified  for  the  most  part  in  terms  of  the  three  circuit  constants,  re- 
sistance, capacitance,  and  inductance. 

The  frequencies  of  interest  in  communication  circuits  range  from  a  few  cycles  per  second 
to  the  super  high  frequencies  used  in  radio  transmission,  whereas  the  frequency  range  of 
major  interest  from  the  standpoint  of  the  measurement  of  primary  circuit  constants  is 
from  about  30  cycles  to  about  100  megacycles. 


6.  MEASUREMENT  OF  CURRENT 

The  values  of  power  of  interest  have  a  lower  limit  of  about  10  ~16  watt  determined  by 
the  necessity  of  keeping  a  level  above  resistance  noise,  and  an  upper  limit  of  about  1  watt 
in  wire  transmission,  determined  by  the  necessity  of  avoiding  excessive  cross-talk  to  other 
circuits.  The  upper  limit,  of  course,  is  considerably  higher  in  certain  applications  such 
as  radio  transmitters.  The  currents  corresponding  to  these  power  limits  for  the  impedance 
ranges  encountered  range  from  about  0.1  amp  down  to  fractions  of  a  microampere.  The 
three  features  most  desired  in  an  instrument  to  measure  such  currents  are :  an  impedance 
practically  non-reactive  and  independent  of  frequency  and  current  level,  a  method  of 
operation  which  furnishes  effective  values,  and  high  ratio  of  response  to  input  power. 

TYPES  OF  INSTRUMENT.  The  dynamometer  type  and  the  magnetic-vane  type 
measure  effective  values  of  current  but  require  high  input  power.  They  are  seldom  used 
except  at  power  frequencies.  Practically  all  a-c  measurements  of  current  use  as  an  in- 
dicator a  d-c  instrument  of  the  moving  coil-permanent  magnet  type.  The  problem  then 
reduces  to  a  means  of  transforming  the  alternating  current  to  direct  current. 

THE  THERMOCOUPLE  TYPE.  This  has  an  impedance  which  is  practically  a  pure 
resistance  at  all  frequencies,  and  it  measures  effective  values  of  current.  It  is  the  most 
accurate  method  for  the  measurement  of  small  currents  because  it  can  be  calibrated  with 
reversed  direct  current.  Its  principal  disadvantages  are  that,  irrespective  of  the  current 
measured,  the  output  power  is  low,  about  1  microwatt,  and  it  will  not  stand  heavy  over- 
loads. Enclosing  the  couple  in  a  vacuum  improves  its  speed  and  sensitivity  and  reduces 
temperature  errors.  The  usual  type  has  a  heater  in  direct  contact  with  the  junction. 
This  gives  a  maximum  sensitivity  and  speed  and  is  satisfactory  for  all  but  the  highest 
frequencies.  At  very  high  frequencies  the  direct  contact  between  the  a-c  and  d-c  circuits 
introduces  objectionable  couplings  between  the  instrument  and  other  parts  of  the  circuit. 
A  type  having  the  couple  insulated  from  the  heater  is  to  be  preferred  above  about  a  million 
cycles  in  spite  of  a  somewhat  slower  speed  and  slightly  lower  efficiency. 

Typical  vacuum  couples  of  either  the  insulated  or  contact  type  have  a  couple  resistance 
of  about  10  ohms  and  are  designed  to  work  into  a  microammeter  having  a  resistance  of 
about  the  same  value  and  a  full-scale  deflection  of  200  to  300  jua.  Table  1  gives  essential 
information  for  instruments  of  this  type  having  various  ranges.  Their  accuracy  is  usually 

Table  1.    Typical  Ranges  and  Resistances  of  Thermocouple  Instruments 


Range, 
milli- 
amperes 

Input 
Resistance, 
ohms 

Range, 
milli- 
amperes 

Input 
Resistance, 
ohms 

Range, 
milli- 
amperes 

Input 
Resistance, 
ohms 

250 
100 
50 

0.5 
2.0 
4.0 

25 
10 
7 

10 
35 
50 

5.0 
2.0 
1.5 

100 
600 
1000 

MEASUREMENT  OF  VOLTAGE  11-17 

about  1  per  cent  of  full-scale  reading  when  used  with  the  couple  with  which  they  have  been 
calibrated. 

Multirange  milliam meters  are  made  with  self-contained  shunts.  The  accuracy  of  such 
instruments  at  high  frequencies  is  usually  limited  by  the  shunts. 

Contact  Rectifier  Type.  Contact  rectifiers  used  in  conjunction  with  d-c  meters  are  the 
most  rugged  and  efficient  of  the  various  low  current  instruments.  The  most  common  type 
uses  the  copper  vs.  copper-oxide  disk.  These  instruments  have  the  disadvantage,  common 
to  all  rectifiers,  that  they  introduce  modulation  due  to  instantaneous  variation  of  the  input 
resistance  with  current.  This  may  be  reduced  considerably  by  the  common  arrangement 
of  four  units  in  bridge  form,  or  an  equivalent  arrangement  with  two  units  and  a  center 
tapped  transformer.  They  usually  measure  more  nearly  average  than  effective  values, 
thus  giving  a  wave-shape  error.  Their  principal  disadvantages  lie  in  the  variation  of  the 
rectification  properties  with  time  and  temperature  and  the  fact  that  the  input  impedance 
varies  with  current  and  has  a  large  capacitive  component  which  acts  as  a  shunt  across  the 
rectifier  and  renders  their  use  above  about  10,000  cycles  of  little  value,  unless  some  com- 
pensation is  provided.  With  proper  compensation  their  frequency  range  may  be  extended 
considerably.  For  higher  frequencies  silicon  or  germanium  crystal  rectifiers  may  be  used. 

Since  different  ranges  can  be  obtained  by  changing  either  the  size  of  the  disks  or  the 
sensitivity  of  the  d-c  meter,  a  very  wide  current  range  is  available.  It  is  possible  to  obtain 
meters  with  a  full  scale  as  low  as  100  ^ta,  but  their  resistance  is  high,  about  3000  ohms. 
This  makes  them  suitable  for  voltmeters,  which  can  be  obtained  with  ranges  as  low  as 
0.5  volt  with  a  resistance  of  3000  ohms  per  volt. 

The  accuracy  of  this  type  of  instrument  is  usually  limited  to  about  5  per  cent  of  full 
scale,  and  errors  in  excess  of  this  may  occur  with  extremes  of  temperature  and  wave  shape. 

Two-element  vacuum-tube  rectifiers  have  characteristics  somewhat  similar  to  contact 
rectifiers.  They  are  satisfactory  up  to  much  higher  frequencies  but  have  the  disadvantage 
of  requiring  auxiliary  power.  They  are  discussed  further  under  voltmeters. 

MEASUREMENT  OF  HIGH  CURRENT  VALUES.  These  measurements  may  be 
made  by  means  of  low-range  meters  in  conjunction  with  current  transformers  or  shunts. 
Shunts  are  preferred  for  high  frequencies.  They  should  be  designed  to  have  impedance 
characteristics  similar  to  those  of  the  meter,  particularly  when  used  at  high  frequencies, 
so  that  the  current  will  divide  in  the  same  proportion  at  all  frequencies. 

MEASUREMENT  OF  LOW  CURRENT  VALUES.  Measurement  of  current  values 
below  the  range  of  even  the  most  sensitive  instruments  can  be  made  after  amplification  by 
comparison  with  a  known  current  obtained  by  attenuating  a  measured  current  a  definite 
amount. 

7.  MEASUEEMENT  OF  VOLTAGE 

With  the  high  impedances  almost  inevitable  in  an  instrument  for  measuring  low  current 
values,  there  is  no  marked  distinction  between  the  measurement  of  current  and  voltage; 
thus  all  the  methods  discussed  for  current  measurement  will  also  measure  voltage.  How- 
ever, care  must  be  taken  to  avoid  errors  due  to  impedance  change  caused  by  series  reactance 
in  the  leads  at  high  frequencies.  Static  voltmeters  are  also  common.  They  measure 
effective  values,  but  their  low  sensitivity  and  high  input  capacitance  limit  their  use  to 
comparatively  high  voltages  and  low  frequencies.  In  addition  the  following  are  available. 

VACUUM-TUBE  RECTIFIERS.  This  type  provides  the  most  satisfactory  means  of 
voltage  measurement.  Most  vacuum-tube  voltmeters  employ  diode  rectifiers  with  high- 
resistance  loads,  rather  than  triodes,  since  the  indication  is  less  dependent  on  the  tube 
characteristics,  and  because  plate-voltage  variations,  which  are  a  source  of  error  in  the 
triode,  are  eliminated, 

To  measure  low  voltages  the  rectifier  is  either  preceded  by  an  a-c  amplifier  or  followed 
by  a  d-c  amplifier.  The  frequency  range  and  sensitivity  of  the  former  are  limited  by  the 
problem  of  obtaining  broad-band  stabilized  amplification.  Instruments  are  available  in 
multiple-range  models  with  sensitivities  in  the  order  of  1  millivolt,  covering  the  frequency 
range  of  20  cycles  to  5  megacycles.  The  input  impedance  is  in  the  order  of  0.5  megohm 
shunted  by  a  few  micro-microfarads.  Indicated  voltages  are  usually  average  values. 

The  sensitivity  of  diode  voltmeters  is  limited  by  the  diode  characteristic,  and  the 
frequency  is  limited  principally  by  lead  inductance.  To  reduce  this  error,  the  diode  is 
usually  encased  in  a  small  probe  connected  to  the  set  by  a  cable.  Instruments  are  available 
in  multiple-scale  models  with  sensitivities  of  0.1  volt  covering  the  frequency  range  of  20 
cycles  to  100  megacycles.  Input  impedance  is  usually  1  megohm  or  more  at  low  frequency 
but  is  limited  at  high  frequency  by  the  capacitance  and  conductance  of  the  diode,  which 
may  be  as  high  as  7  wi  and  10  micromhos  at  10  megacycles.  Indicated  voltages  are  usually 
half  wave  peak  values. 


11-18  ELECTRICAL  MEASUREMENTS 

The  accuracy  of  vacuum-tube  voltmeters  is  2  to  5  per  cent,  depending  on  wave  shape  and 
frequency. 

For  frequencies  above  the  range  of  the  diode,  materials  of  high  temperature  coefficient 
of  resistance  are  used  in  which  the  voltage  is  determined  by  the  resistance  change  due  to 
heating.  Owing  to  their  small  size  they  may  be  made  independent  of  frequency  up  to 
several  thousand  megacycles. 

8.  RESISTANCE  STANDARDS 

For  the  accurate  measurement  of  circuit  constants,  precision  standards  are  required. 
They  form  the  most  important  part  of  the  measuring  circuits  and  therefore  require  full 
consideration.  They  consist  of  resistance,  capacitance,  and  inductance.  The  following 
discussion  is  limited  to  their  use  as  standards.  Further  general  information  can  be  found 
in  Section  3. 

Resistance  standards  are  usually  wire  wound,  but  high  resistances  are  also  made  by 
coating  a  thin  film,  usually  of  carbon,  on  an  insulating  form.  These  films  have  a  high  tem- 
perature coefficient  of  resistance,  about  —  0.03  per  cent  per  degree  centigrade  for  carbon, 
and  cannot  be  adjusted  to  very  close  limits.  They  are  fairly  stable  with  time  if  hermetically 
sealed,  and  the  change  with  frequency  is  small  for  values  below  10,000  ohms.  They  are 
cheaper  than  wire  wound,  especially  for  high  resistance  values. 

The  types  of  wire  used  most  generally  for  resistance  standards  are  the  copper-nickel 
alloy  Constantan  and  the  copper-nickel-manganese  alloy  Manganin.  Both  materials  have 
very  low  temperature  coefficients,  the  variation  in  resistance  over  the  temperature  range 
of  ordinary  use  being  well  below  0.005  per  cent.  Constantan  is  more  easily  soldered,  and 
its  resistance  is  more  stable  with  time.  Manganin  is  preferred  for  d-c  standards  on  account 
of  its  lower  thermoelectric  power  to  copper,  but  this  is  a  minor  consideration  for  a-c  use. 
Both  alloys  have  a  specific  resistance  of  about  30  times  that  of  copper.  Other  alloys  con- 
taining chromium  or  iron  have  higher  specific  resistances  but  are  not  generally  used  on 
account  of  their  higher  permeabilities  and  larger  temperature  coefficients. 

EFFECT  OF  HUMIDITY.  Humidity  affects  the  distributed  capacitance  of  resistance 
windings  unless  they  are  sealed  or  impregnated.  The  impregnating  material  is  usually  a 
thin  solution  of  shellac  or  a  wax,  such  as  paraffin.  Impregnation  also  serves  to  prevent 
change  in  shape  of  the  support  or  spool,  but  it  is  desirable  to  use  as  a  support  a  material 
impervious  to  moisture,  such  as  a  suitable  plastic,  ceramic,  or  glass.  The  woven  type  of 
wire  standard  is  the  most  independent  of  changes  in  the  form. 

PHASE  ANGLE.  The  phase  angle  should  be  small.  It  is  due  to  inductance,  distributed 
capacitance,  and  capacitance  between  the  terminals  which  is  not  due  to  the  presence  of  the 
winding.  Inductance  can  be  reduced  by  reducing  the  size  of  wire  and  by  winding  the  coil 
BO  that  adjacent  turns  have  opposite  directions  of  winding  and  are  spaced  as  close  together 
as  possible.  Minimum  spacing  is  determined  by  the  thickness  of  insulation.  Therefore, 
for  a  given  type  of  wire  and  insulation  there  is  a  definite  mininiuni  obtainable  ratio  of 
inductance  to  resistance  for  any  given  wire  size,  this  value  being  reduced  as  the  wire  size 
is  reduced.  Distributed  capacitance  can  be  reduced  by  winding  so  that  turns  which  are 
adj  acent  physically  are  consecutive  electrically.  By  this  means  the  distributed  capacitance 
can  be  reduced  below  the  remaining  capacitance  between  the  terminals.  Both  types  of 
capacitance  can  be  decreased  by  reducing  the  physical  size,  that  is,  by  reducing  the  wire 
size.  Capacitance  and  inductance,  when  both  are  present  in  a  resistance,  have  a  com- 
pensating effect.  For  the  small  reactances  present  in  resistance  standards,  the  compensa- 
tion will  be  almost  perfect  at  all  frequencies  if  the  relation  L  =  Cr2  holds. 

The  reactance  of  carbon  film  type  standards  may  be  considered  due  entirely  to  capaci- 
tance between  terminals.  This  may  be  held  to  about  1  ppf. . 

VARIATION  WITH  FREQUENCY.  Resistance  change  with  frequency  is  due  prin- 
cipally to  the  presence  of  skin  effect  and  of  residual  reactances.  Skin  effect  may  be  reduced 
by  using  a  small  size  of  wire  and  reducing  the  inductance  of  the  winding  by  suitable  choice 
of  winding  type.  For  non-inductive  windings,  the  increment  due  to  skin  effect  may  be 
taken  as  less  than  0.1  per  cent  for  No.  28  B.&S.  gage  wire  at  1  megacycle,  and  less  than  1 
per  cent  for  No.  30  wire  at  10  megacycles.  Where  power  considerations  do  not  enter, 
sizes  smaller  than  these  are  generally  used  in  order  to  reduce  the  phase  angle  as  already 
discussed,  and  skin  effect  is  therefore  usually  negligible. 

Reactance  is  a  more  serious  cause  of  resistance  variation  with  frequency.  If  a  resistance 
has  inductance  in  series  with  it,  the  effective  resistance  component  of  the  combination 
eonsidef  ed  as  a  parallel  circuit  will  be  increased  to  the  value  r  +  (bPLP/r)  -  If  the  resistance 
feas  capacitance  in  parallel  with  it,  the  effective  resistance  of  the  combination  considered 
as  a  series  circuit  will  be  decreased  to  the  value  r  —  o^CV.  When  both  inductance  and 


RESISTANCE  STANDARDS 


11-19 


capacitance  are  present,  the  value  of  resistance  is  a  function  of  both,  and,  if  the 
inductance  and  capacitance  are  in  such  proportions  as  to  give  it  a  zero  phase  angle,  its 
value  will  be  increased  to  r  -f-  (<*rL?/r).  For  high  resistances  at  high  frequencies,  the 
dielectric  loss  of  the  associated  capacitance  affects  the  resistance  value. 

TYPES  OF  WINDINGS.  The  following  windings  are  suitable  for  resistance  standards 
having  values  above  about  50  ohms,  for  which  cases  the  reduction  of  both  inductance  and 
capacitance  is  important: 

The  Curtis  winding  made  by  winding  one  turn  around  a  spool,  then  passing  the  wire 
through  an  axial  slot  and  winding  the  next  turn  in  the  opposite  direction.  A  flat  slotted 
form  or  card  is  an  improvement  over  a  round  spool. 

The  inductive  winding  on  a  thin  flat  card. 

The  woven  type  in  which  the  warp  consists  of  silk  or  cotton  threads  and  the  weft  is  the 
resistance  wire. 

The  Ayrton-Perry  type  consisting  of  two  parallel  opposed  windings  either  in  a  single 
layer,  in  which  case  they  must  cross  at  every  turn,  or  one  layer  wound  over  the  other.  The 
single  layer  has  a  little  more  capacitance  and  a  little  less  inductance  than  the  two  layers. 

For  resistance  below  50  ohms  where  capacitance  has  less  effect,  windings  having  greater 
distributed  capacitance  such  as  the  bifilar,  and  the  reversed  layer  type  may  be  used. 

A  well-made  unit  using  No.  36  B.&S.  gage  wire  may  be  wound  to  have  a  ratio  of  react- 
ance to  resistance,  due  to  inductance,  of  about  0.1  at  I  megacycle,  and,  using  No.  44  wire, 
to  have  a  ratio  of  about  0.2  at  10  megacycles.  If  the  capacitance  is  such  as  to  give  zero 
phase  angle,  which  it  may  for  windings  of  about  1000  ohms,  the  corresponding  resistance 
increment  will  be  about  1  and  4  per  cent  respectively.  If  the  capacitance  value  is  not  such 
as  to  give  zero  phase  angle,  the  effective  shunt  resistance  will  be  different  from  the  effective 
series  resistance. 

Table  2  gives  representative  values  of  inductance  and  capacitance  for  standards  of 
several  values  wound  with  various  sizes  of  wire. 

Table  2.    Representative  Resistance  Windings 

Inductance  and  capacitance  for  various  values  and  wire  sizes 


Resistance, 
ohms 

Type  of  Winding 

Insulation  and 
Wire  Size 
B.&S.  Gage 

Residuals 

Net  Residual 

L,fb 

c, 

w*f 

I/,  ;ih 

C'  =  £'/r2, 
fifd 

0.1 
1 
10 
100 
100 
100 
100 
1,000 
1,000 
1,000 
1,000 
10,000 
10,000 
10,000 

Bifilar,  tape 
Bifilar,  tape 
Bifilar 
Bifilar 
Curtis  (card) 
Ayrton-Perry  (2  layers) 
Woven 
Reversed  layer 
Ayrton-Perry  (2  layers) 
Woven 
Curtis  (card) 
Reversed  layer 
Ayrton-Perry  (1  layer) 
Woven 

0.004  in.  X  0.1  25  in. 
0.004  in.  X  0.063  in. 
33  DSC 
S6DSC 
40  DSC 
40  BE 
40  BE 
39  DSC 
44  BE 
44  DSC 
40  DSC 
42  DSC 
44  DSC 
44  DSC 

0.01 
0.04 
0.16 
0.8 
0.4 
0.35 
0.30 
4.2 
2.5 
2.0 
4.0 
30 
25 
20 



0.01 
0.04 



10 
80 
0.5 
1 
! 
24 
1 
0.5 
1.5 
20 
2 
0.5 

0.16 
0 
0.4 
0.34 
0.29 
-20 
1.5 
1.5 
2.5 

0 
-40 
-34 
-29 
20 
-1.5 
-1.5 
-2.5 
20 
1.8 
0.3 

VARIABLE  STANDARDS.  Variable  standards  are  used  in  precision  measurements 
principally  for  covering  the  range  between  the  smallest  steps  of  the  adjustable  standards. 
They  are  usually  wire  wound  and  of  low  range.  The  simple  series  type  has  the  abjection 
that  the  contact  resistance  is  usually  an  appreciable  part  of  the  total  resistance  of  the 
standard.  Preferred  methods  are  to  use  the  slide  wire  in  a  circuit  as  a  potentiometer,  in 
which  case  the  contact  resistance  is  not  in  the  measuring  circuit,  or  to  use  a  shunted  type 
of  slide  wire,  in  which  case  the  resistance  of  the  slide  wire  itself  is  higher  than  the  range 
of  the  combination,  thus  reducing  the  effect  of  the  contact  resistance.  Inductance  con- 
stitutes the  principal  frequency  limitation.  The  shunted  type  has  lower  inductance  than 
the  potentiometer  type  but  does  not  have  a  linear  scale  unless  used  as  a  potentiometer. 
Compensation  for  inductance  in  single-turn  slide  wires  may  be  made  by  substituting  an 
equivalent  length  of  copper  wire  for  the  resistance  wire  removed  from  the  circuit.  Com- 
pensation in  the  wound  type  is  not  generally  made  but  can  be  effected  in  special  eases  by 
winding  the  slide  wire  non-inductively. 

High-resistance  variable  standards  are  sometimes  used  in  shunt  connection  as  low-grange 
conductance  standards.  These  are  commonly  a  composition  type,  particularly  when  used 
at  high  frequencies.  They  are  not  very  stable  standards. 


11-20 


ELECTRICAL  MEASUREMENTS 


Resistor 


ADJUSTABLE  STANDARDS.  Adjustable  standards  consist  of  a  number  of  switch 
assemblies,  usually  1  to  6,  each  containing  resistance  units  arranged  to  give  values  in  steps 
from  1  to  10.  The  switch  and  wiring  add  both  inductance  and  capacitance.  These  may 

be  reduced  by  reducing  the  switch  size 
and  the  amount  of  dielectric  material  in 
it.  These  requirements  are  met  very 
well  by  the  wafer-type  switch.  The 
small  size  introduces  some  difficulties  in 
obtaining  satisfactory  low  and  stable 
contact  resistance,  but  the  introduction 
of  silver  contacts  reduces  this  difficulty. 
Their  principal  objection  is  their  short 
life.  They  have  considerable  flexibility 
due  to  the  availability  of  multiple  decks. 
The  usual  arrangement  is  to  connect  10 
equal  resistances  in  series  and  to  short- 
circuit  those  not  in  use  by  means  of  the 
brush.  The  ground  or  low-potential  side 
of  the  circuit  should  be  connected  to  the 
brush  to  minimize  the  effect  of  capaci- 
tance. The  inductance  may  be  compen- 
sated for  by  using  an  additional  10  studs 
as  shown  in  Fig.  1,  to  insert  inductance 
equal  to  the  inductance  of  the  resistance 
and  wiring  which  is  removed,  thus  keep- 
ing the  total  circuit  inductance  constant 
for  all  switch  settings  at  the  cost  of  a  slight  increase  of  the  zero  inductance.  This  arrange- 
ment uses  2  decks  of  a  wafer  switch. 

A  method  which  reduces  both  capacitance  and  inductance  to  a  minimum,  and  is  there- 
fore suitable  for  both  high  and  low  resist- 
ances, is  shown  in  Fig.  2.  A  20-stud  switch  is 
so  arranged  that  the  drum  supporting  the 
units  and  studs  rotates,  the  brushes  remain- 
ing stationary.  Each  unit  in  a  decade  has  a 
resistance  equal  to  the  total  value  required  for 
that  particular  setting.  There  is  no  switch 
wiring  and  there  are  no  coils  connected  when 
not  in  use.  The  short-circuit  connection  for 
the  zero  setting  may  consist  of  a  copper  strap 
equal  in  inductance  to  the  mean  value  of  the 
units.  The  same  arrangement  with  station- 
ary studs  and  moving  brushes  is  almost  as 
satisfactory  and  allows  the  use  of  a  2-deck 
wafer  switch.  The  principal  disadvantage  is 
the  necessity  for  10  different  values  for  the 
resistance  units. 

For  high  resistances  there  are  advantages 
in  arranging  decades  to  read  conductance. 
This  means  connecting  them  in  parallel. 


Compensating 

Inductor  Brushes 

FIG.   1.     Decade  Resistance  Standard,   Compensated 
for  Inductance 


100  ohm  decade 


FIG.  2.    Rotor  Type  Decade  Resistance  Standard 


Four  units  of  value  1,  2,  3,  4  can  be  connected  in  various  parallel  combinations.    The 
switching  arrangement  is  the  same  as  that  for  switching  capacitors,  shown  in  Fig.  5A. 


9.  CAPACITANCE  STANDARDS 

The  principal  requirements  for  a  capacitance  standard  are  a  capacitance  independent 
of  time,  humidity,  temperature,  voltage  and  frequency;  a  low  power  factor;  and  small 
size.  The  degree  to  which  these  requirements  can  be  met  depends  principally  on  the  choice 
of  dielectric.  All  capacitors  using  solid  or  liquid  dielectrics  have  losses  when  subjected  to 
alternating  voltages.  The  equivalent  circuit  of  such  a  capacitor  may  be  indicated  as  shown 
in  Fig.  3  as  a  perfect  capacitance  C  either  with  a  resistance  rs  in  series  with  it,  or  with  a  con- 
ductance g  in  parallel  with  it.  For  these  networks  to  be  equivalent  the  two  capacitances 
are  not  identical,  but  for  the  low  power  factors  of  standards,  the  difference  is  negligible. 
i.  Practically  the  only  dielectrics  used  for  capacitance  standards  are  mica  or  air.  Many 
other  solid  dielectrics  are  used  for  capacitors.  None  of  them  combines  all  the  advantages 


CAPACITANCE  STANDARDS 


11-21 


a>  CB> 

FIG.  3.     Alternate  Equivalents  for  an  Im- 
perfect Capacitor.     A,  parallel  equivalent; 
B,  series  equivalent. 


of  mica,  although  polystyrene  is  equal  or  superior  to  mica  in  practically  all  respects  except 
temperature  coefficient  of  capacitance. 

Mica  capacitors  may  be  made  of  interleaved  sheets  of  mica  and  foil  such  as  copper,  tin, 
or  aluminum,  or  by  depositing  a  film  of  metal,  usually  silver,  directly  on  the  mica.    The 
last  has  the  advantage  of  eliminating  any  air  film  between  the  dielectric  and  the  metal, 
making  the  capacitance  less  dependent  on  the 
mechanical  stability  of  the  assembly.  c 

Air  capacitors  require  solid  dielectric  in  their  j 1  I r  c 

construction.   Their  performance  depends  consid-     '  l         •  j r* 

erably  on  their  design  as  well  as  on  the  dielectric.  1  I       VW\> 

STABILITY  WITH  TIME  AND  HUMIDITY.  L/ys/^J 

All  capacitors  change  slightly  with  age,  owing  to  "  —  — 

structural  changes  after  assembly;  good  mica 
capacitors  usually  increase  less  than  0.05  per  cent 
with  age.  Humidity  has  a  serious  effect  unless 
the  capacitors  are  protected.  They  are  usually 
dried  and  hermetically  sealed,  either  in  molded 

plastic  or  in  a  can.    In  addition  they  may  be  vacuum-impregnated,  usually  with  paraffin 
or  Superla  wax.    Well-made  air  capacitors  are  less  subject  to  these  changes. 

VARIATION  WITH  TEMPERATURE.  The  change  in  capacitance  with,  temperature 
of  mica  capacitors  is  due  partly  to  the  dielectric  material  but  is  also  affected  considerably 
by  the  mechanical  design.  Unimpregnated  capacitors  usually  have  a  positive  linear 
change,  which  is  due  primarily  to  changes  in  the  physical  dimensions.  In  impregnated 
capacitors,  additional  changes  occur  owing  to  the  effect  of  the  wax,  which  has  a  negative 
temperature  coefficient  that  increases  with  temperature.  As  a  result,  the  temperature 
coefficient  of  the  capacitor  depends  on  the  amount  of  wax  and  may  be  made  extremely 
small  over  a  limited  temperature  range.  Standards  can  be  obtained  which  vary  less  than 
0.05  per  cent  from  10  deg  cent  to  35  deg  cent.  The  shunt  conductance  of  all  mica  capacitors 
increases  with  temperature.  The  amount  varies  considerably,  but  an  average  value  may 
be  taken  as  2  per  cent  per  deg  cent.  For  air  capacitors  capacitance  change  is  due  partly 
to  the  insulating  supports  but  principally  to  change  in  the  plate  area  and  spacing  caused 
by  changes  in  the  dimensions  of  the  parts.  It  may  be  reduced  by  the  choice  of  materials 
of  low  temperature  coefficient  of  expansion  and  by  avoiding  distortion  due  to  unequal 
expansion  of  the  parts. 

VARIATION  WITH  FREQUENCY.  Variations  in  capacitance  with  frequency  of  mica 
standards  are  caused  by  variations  in  the  dielectric  constant,  and  at  high  frequencies  by 
the  effect  of  inductance  in  the  leads.  The  former  causes  a  slight  decrease  in  capacitance 
with  increasing  frequency,  the  change  being  approximately  logarithmic  with  respect  to 
frequency.  The  latter  causes  an  increase  in  capacitance  equal  to  oo^LC2.  Lead  inductance 
in  a  good  standard  may  be  held  below  0.05  juh.  Air  capacitors  change  less,  because  of 
the  smaller  amount  of  dielectric  material  and  because  their  comparatively  low  capacitance 
is  affected  by  lead  inductance  only  at  higher  frequencies. 

An  imperfect  capacitor  may  be  represented  as  shown  in  Fig.  ZA  or  Fig.  3B.  Metallic 
losses  constitute  a  series  resistance  which  is  independent  of  frequency,  except  for  skin 
effect.  Leakage  constitutes  a  shunt  conductance  independent  of  frequency.  Dielectric 
loss  is  of  such  a  nature  that  the  loss  per  cycle  is  proportional  to  capacitance  and  approxi- 
mately independent  of  frequency.  In  other  words,  this  loss  is  proportional  to  &C.  If 

represented  by  Fig.  3A,  g  will  be 

TaWe  3.     Loss  Variation  with  Frequency  and          directly  proportional  to  frequency, 
Capacitance  and  if  represented  by  Fig.  3£,  rs 

will  be  inversely  proportional  to 
frequency. 

In  general,  four  different  quan- 
tities are  in  common  use  for  indi- 
cating the  quality  of  a  capacitor 
from  the  standpoint  of  loss.  These 
are  series  resistance  r,  shunt  con- 
ductance g,  power  factor  which 
when  small  is  equal  to  the  dissipa- 
tion factor  coCV  or  g/&C,  and  the 
reciprocal,  which  is  commonly  termed  Q.  Table  3  indicates  how  these  quantities  vary 
with  frequency  and  capacitance. 

In  this  table  D  is  a  fundamental  property  of  the  dielectric  and  is  approximately  inde- 
pendent of  the  frequency.  For  good  mica  it  is  as  low  as  0.0001  and  decreases  slightly  with 
increasing  frequency. 


Source  of  Loss 

r 

Q 

P.F. 

g 

Series  resistance  r,  

ra 

I 

r&C 

r^-C* 

Dielectric,  D  >   . 

D 

1 

T\ 

«C 

D 

D-c  leakage  go 

So 

uC 

SO 

go 

03    C2 

£0 

wC 

11-22 


ELECTRICAL  MEASUREMENTS 


Low  power  factor  or  high  Q  in  a  standard  is  obtained  by  choice  of  mica  of  low  dielectric 
loss,  by  keeping  lead  resistance  low,  and  by  eliminating  moisture  and  impurities  in  assem- 
bly. Figure  4  shows  curves  of  power  factor  and  Q  in  terms  of  frequency  for  representative 
standards.  The  increase  in  Q  with  frequency  is  a  characteristic  of  the  dielectric.  The 

decrease  at  high  frequencies  is  due 
to  effective  series  resistance.  The 
larger  the  value  of  capacitance,  the 
lower  is  the  frequency  at  which  the 
maximum,  occurs.  Small  capaci- 
tance values  usually  have  greater 
dielectric  loss  because  of  the  greater 
influence  of  the  sealing  material. 
They  also  have  greater  effective 
series  resistance,  owing,  in  part  at 
least,  to  the  smaller  number  of 
sheets  in  parallel. 

The  Q  of  air  standards  is  usually 
higher  than  that  of  mica  on  ac- 
count of  the  reduced  dielectric  ma- 
terial and  the  reduced  effect  of 
series  resistance  due  to  the  smaller 
capacitance. 


30  100 

Frequency  in  kilocycles 


300 


1000 


PIG.  4.     Variation  with  Frequency  of  Q  of  Typical  Mica 
Capacitor  Standard 


VARIABLE  STANDARDS.  Variable  standards  are  practically  always  air  capacitors. 
They  may  be  considered  as  two  capacitances  in  parallel,  one  consisting  of  the  Tnirnmum 
capacitance  which  includes  the  dielectric  material,  and  the  other  consisting  of  the  variable 
part.  The  former  may  be  considered  as  a  fixed  capacitance  shunted  by  a  conductance 
representing  the  dielectric  loss,  which  is  not  a  function  of  setting.  The  latter  may  be 
considered  as  a  loss-free  variable  capacitance  in  parallel  with  the  former.  They  may  be 
used,  therefore,  in  measuring  circuits,  to  obtain  loss-free  changes  hi  capacitance.  On  the 
other  hand,  considering  the  standard  as  a  whole,  the  power  factor  increases  with  decrease 
in  setting  and  may  be  greater  than  that  of  a  mica  capacitor  at  minimum  setting  unless  a 
very  low  loss  dielectric  is  used. 

ADJUSTABLE  STANDARDS.  These  are  usually  in  decade  form,  but,  owing  to  the 
cost  of  good  standards,  only  4  units  are  gener- 
ally used,  the  switch  connecting  them  in  the 
parallel  combinations  necessary  to  obtain  steps 
from  1  to  10.  Practically  all  such  switches  re- 
duce to  the  principle  of  a  brush  for  each  unit 
contacting  successively  10  studs  which  are 
wared  so  as  to  connect  the  unit  in  circuit  at  the 
positions  required  to  give  the  desired  combina- 
tions. This  may  be  most  readily  done  with 
minimum  wiring  by  using  cams.  The  method 
is  indicated  by  the  developed  cam  arrangement 
shown  in  Fig.  5.4. .  By  using  2  brushes  on  dif- 
ferent diameters  a  single  cam  may  be  cut  to 
connect  2  units.  Wafer  switches  can.  be  used 
requiring  only  2  cams  and  5  brushes  mounted 

on  a  single  deck  to  perform  the  whole  switching  o    1    2    3   4    5   6 

sequence.    A  similar  switch  may  be  mounted  on  A  senes  connection 

the   same  shaft  with  cams  cut  inversely  as    fiq-   5.    Cam  Type  of  Decade  Switch  for 
,  •     -n-      EI>  i.    *  71  T  *  j  *     •         Senes  or  Parallel  Connection  of  4  Standards, 

shown  in  Fig.  5B  but  parallel-connected  to  an-    A,  parallel  connection;  B,  series  connection, 
sert  capacitances  in  place  of  those  removed 

to  compensate  for  the  capacitance  of  the  switch  or  for  that  of  the  units  in  excess  of 
nominal  values. 

10.  INDUCTANCE  STANDARDS 

The  requirements  for  inductance  standards  are  similar  to  those  for  capacitance  stand- 
ards, namely,  high  stability  of  inductance  under  all  operating  conditions,  low  power  factor 
or  high  Qt  and  small  size.  Coils  cannot  be  built  to  meet  these  requirements  as  well  as 
capacitors,  and  for  this  reason  they  are  not  used  to  the  same  extent  as  standards.  For 
precise  standards  no  magnetic  material  is  sufficiently  stable,  and  air-core  coils  are  almost 
aNrays  used.  They  are  wound  either  as  toroids  or  as  solenoids.  The  toroids  have  tees 
external  field  but  are  larger  for  the  same  performance.  External  field  is  objectionable 


MEASUREMENT  OF  RESISTANCE  11-23 

since  it  not  only  causes  errors  in  measurement  due  to  coupling  to  the  rest  of  the  circuit 
but  the  inductance  of  the  standard  itself  is  affected  also.  Shielding  is  generally  used  con- 
sisting of  magnetic  material  at  low  frequencies  and  low-resistance  material  such  as  copper 
at  high  frequencies. 

STABILITY  WITH  TIME  AND  HUMIDITY.  Coils  may  be  constructed  to  have 
negligible  change  in  inductance  and  resistance  with  time.  They  are  usually  sealed  and 
in  addition  may  be  impregnated  to  eliminate  humidity  effects. 

VARIATION  WITH  TEMPERATURE.  Change  in  inductance  is  determined  prin- 
cipally by  mechanical  changes  in  the  form  and  in  the  impregnating  material  when  present. 
By  choice  of  materials  of  low  expansion  these  changes  can  be  held  to  as  low  as  0.001  per 
cent  per  deg  cent.  A  single-layer  solenoid  wound  under  tension  on  a  ceramic  form  may 
be  made  with  a  temperature  coefficient  of  less  than  2  parts  per  million  per  deg  cent. 
Change  of  resistance  is  due  principally  to  change  in  the  resistivity  of  the  copper  winding 
and  to  change  in  eddy-current  loss  which  in  turn  is  due  to  the  change  in  resistivity.  These 
changes  are  of  opposite  sign.  The  net  change  is  a  function  of  frequency  but  is  usually 
positive. 

VARIATION  WITH  FREQUENCY.  Variations  of  inductance  and  effective  resistance 
with  frequency  are  due  principally  to  eddy  currents  and  distributed  capacitance.  The 
former  results  in  an  increment  of  resistance  which  is  proportional  to  the  square  of  the 
frequency.  It  can  be  reduced  by  using  stranded  wire  and  single-layer  windings.  The 
effect  on  the  inductance  is  usually  negligible  compared  with  other  effects.  Distributed 
capacitance  has  the  effect  of  increasing  both  inductance  and  effective  resistance.  It  is  a 
minimum  for  single-layer  coils.  Where  this  type  is  impractical,  banked  or  sectionalized 
windings  may  be  used.  Capacitance  to  ground  can  be  reduced  appreciably  only  by  reduc- 
ing the  size  of  the  coil,  which  is  at  the  cost  of  Q.  For  frequencies  well  below  the  resonant 
frequency  of  the  coil,  the  increment  in  inductance  due  to  capacitance  C  is  w^C,  and  the 
increment  in  the  effective  resistance  is  2co2LCr.  The  dielectric  conductance  g  associated 
with  the  distributed  capacitance  may  also  be  appreciable.  It  increases  the  resistance  of 
the  coil  by  gcJL*.  If  the  coil  has  no  appreciable  resistance  or  inductance  variation  with 
frequency,  then  the  ratio  of  reactance  to  resistance  or  Q  is  proportional  to  the  frequency. 
On  the  other  hand  the  loss  due  only  to  capacitance  and  eddy  currents  results  in  a  Q  in- 
versely proportional  to  the  frequency.  For  the  actual  case,  the  Q  usually  increases  to^a 
maximum  and  then  decreases.  For  low  frequencies  a  high  Q  can  be  obtained  only  by  means 
of  a  large  size.  This  is  the  principal  disadvantage  of  air-core  standards.  It  is  most  evident 
when  they  must  be  used  for  measuring  magnetic-core  coils.  As  the  frequency  increases 
the  comparison  becomes  more  favorable  to  the  air-core  coils  on  account  of  the  increase  in 
core  loss  and  decrease  in  effective  permeability  of  magnetic  core  coils.  Above  100  kc  this 
disadvantage  is  negligible,  and  air-core  coils  can  be  built  of  sma,ll  size  to  have  values  of  Q 
as  high  as  100  to  200. 

INDUCTOMETERS.  All  inductometers  have  an  appreciable  stray  magnetic  field 
which  may  cause  error  due  to  magnetic  coupling  to  other  parts  of  the  circuit.  They  have 
characteristics  similar  to  fixed  standards  but  usually  have  relatively  greater  capacitance 
and  greater  resistance  increment.  The  inductance  and  resistance  increments  are  a  func- 
tion of  the  setting  as  well  as  of  frequency. 

One  type  consists  of  two  coils  arranged  so  that  the  position  of  the  field  of  one  may  be 
varied  so  as  to  change  their  mutual  inductance.  They  may  be  used  as  variable  mutual 
inductors  or,  by  connecting  the  two  windings  in  series,  as  variable  self-inductors.  Another 
type  consists  of  a  single-layer  rotating  solenoid  with  spaced  bare  wire  turns  and  a  fixed 
brush  riding  on  the  turns.  The  former  type  has  the  advantage  of  no  sliding  contacts,  but 
the  latter  has  better  performance  in  other  respects. 

ADJUSTABLE  STANDARDS.  Coils  may  be  assembled  to  form  adjustable  standards 
similar  to  resistance  and  capacitance  standards.  Because  of  their  large  size,  poorer  char- 
acteristics, and  difficulty  in  compensating  for  capacitances  of  switch  and  wiring  with  series 
connection,  they  are  not  used  extensively.  A  typical  method  of  connecting  4  units  to 
form  a  decade  is  shown  in  Fig.  5.5.  Additional  decks  with  cams  cut  inversely  may  be 
used  to  insert  compensating  resistances  in  place  of  inductances  removed. 

11.  MEASUREMENT  OF  RESISTANCE 

The  method  used  almost  exclusively  for  measuring  circuit  constants  is  the  bridge  method 
in  which  the  unknown  quantity  is  compared  with  a  standard  of  known  value.  For  the 
higher  frequencies,  say,  above  10  megacycles,  the  errors  caused  by  the  presence  of  even 
very  small  stray  inductance  or  capacitance  limit  the  range  and  flexibility  of  the  bridge 


11-24 


ELECTRICAL  MEASUREMENTS 


method,  and  consequently  at  these  frequencies  other  methods  which  involve  less  apparatus 
and  simpler  circuits  are  also  used. 

Resistance  standards  can  be  made  so  that  their  effective  resistance  does  not  differ  from 
the  d-c  value  appreciably  compared  with  the  errors  of  measurement  up  to  the  maximum 
frequencies  used-  Accordingly,  the  measurement  of  resistance  is  usually  made  by  direct 
comparison  with,a  standard  on  a  bridge.  The  simplest  type  is  the  equal  ratio-arm  bridge. 

A  modified  equal  ratio-arm  bridge  useful  for  the  a-c  mea- 
surement of  both  resistance  and  phase  angle  is  shown  in 
Fig.  6.  It  is  adaptable  only  to  the  measurement  of  re- 
sistors having  small  phase  angles,  but,  with  a  given  air 
capacitor  Cb  of  small  range  across  one  ratio  arm,  it  will 
measure  the  reactance  of  resistors  of  any  value  over  a 
wide  frequency  range.  The  equations  of  balance  for 
this  bridge  for  resistors  of  small  phase  angle  are 


/-r 

Cx 


T 
or     Lx 


and 


FIG.  6.    Bridge  Circuit  for  Measure- 
ment of  Resistance  and  Phase  Angle 
of  Resistora 


where  r,  is  the  resistance  of  the  standard. 

rx,  Cx,  and  Lx  are  the  values  for  the  unknown. 
Both  positive  and  negative  reactance  can  be  measured 
by  either  transposing  the  standard  and  unknown  in  the 
bridge  or  by  transferring  the  capacitance  Cb  from  EC 
to  AB.    If  the  standard  has  appreciable  reactance,  cor- 
rection for  it  is  necessary.    For  maximum  accuracy  the 
bridge  should  be  shielded.    A  method  of  shielding  simi- 
lar to  that  shown  in  Fig.  8  is  satisfactory. 
At  very  high  frequencies  a  direct  substitution  in  a  resonant  circuit  as  described  under 
"Measurement  of  Inductance"  may  be  used,  or  a  voltmeter-ammeter  method,  with  a 
vacuum  tube  voltmeter  and  a  thermocouple. 

Calibration  of  standards  for  phase  angle  may  be  made  by  substitution  of  one  resistance 
by  another  having  the  same  physical  configuration  but  a  different  resistance  value,  thus 
obtaining  a  change  in  resistance  without  changing  the  associated  inductance  or  capaci- 
tance. For  low  resistances  this  may  be  done  by  making  identical  standards  of  copper  and 
of  various  resistance  alloys.  For  high  resistances,  carbon  of  different  thickness  may  be 
deposited  on  identical  forms. 

The  measurement  of  effective  resistance  associated  with  inductance  and  capacitance  is 
considered  under  those  headings. 


12.  MEASUREMENT  OF  CAPACITANCE  AND  CONDUCTANCE 

TYPES  OF  CAPACITANCE.  In  practice,  a  capacitor  does  not  usually  consist  of  a 
single  capacitance  between  two  terminals.  There  is,  in  addition,  capacitance  to  ground  or 
to  other  objects.  The  capacitor  is  usually  equivalent  to 
a  three-terminal  network  of  three  capacitances  even  when 
reduced  to  its  simplest  form.  As  a  result,  different  values 
of  capacitance  will  be  obtained,  depending  on  the  condi- 
tions of  measurement.  Referring  to  Fig.  7,  Ci  is  the  ca- 
pacitance to  ground  from  A,  and  C$  is  the  capacitance  to 
ground  from  B.  The  direct  capacitance  between  A  and  B 
is  C;  the  mutual  capacitance  is  C  +  CiC^/(C\  +  Cs) ;  and 
the  grounded  capacitance  is  C  -f-  Ci  or  C  +  Cz,  depending 
on  whether  B  or  A  is  grounded.  Any  or  all  of  these  ca- 
pacitances may  require  measurement. 

In  actual  use,  the  capacitor  is  generally  connected  either 
from  one  side  of  the  system  to  ground,  in  which  event  it 
is  a  grounded  capacitance  which  is  effective,  or  directly 


FIG.  7.     Simplified  Network  of  Ca- 
pacitor Which  Has  Capacitance  to 
Ground 


across  the  system  which  is  balanced  to  ground,  in  which  event  it  is  the  mutual  capacitance 
that  is  effective.  Direct  capacitance  is  required,  in  general,  only  for  special  purposes  or 
where  the  capacitor  is  so  connected  in  service  that  the  three  direct  capacitances  are 
connected  across  different  parts  of  the  circuit  and  must  be  known  individually. 

BRIDGE  METHODS  OF  MEASUREMENT.    Owing  to  the  satisfactory  characteristics 
01  capacitance  standards,  the  measurement  of  capacitance  is  usually  made  by  direct  com- 


MEASUREMENT  OF  CAPACITANCE  AND  CONDUCTANCE       11-25 


parison  with  them.  The  simplest  and  most  accurate  method  is  the  equal  ratio-arm  bridge. 
The  advantages  of  this  type  of  bridge  over  all  other  types  are  that  the  equality  of  the 
ratio  arms  may  be  tested  in  the  bridge  itself  by  means  of  simple  reversal,  and  residual  and 
lead  impedances  may  be  more  readily  balanced  because  of  the  symmetrical  arrangement. 
The  ratio  arms  are  usually  resistances.  Capacitances  have  advantages  for  high  voltages, 
and  inductances,  particularly  when  closely  coupled,  for  high  currents.  For  precise  work, 
especially  at  the  higher  frequencies,  some  method  of  taking  care  of  undesired  capacitances 
between  arms  and  from  the  arms  to  ground  is  necessary,  the  best  method  being  to  shield 
the  bridge.  This  shielding  should  include  at  least  one  transformer.  The  shield  should  be 
complete  enough  to  reduce  the  direct  capacitance  between  windings  below  1  /i;xf-  In 
special  cases  it  should  be  much  better  than  this. 

Capacitance.  Figure  8  shows  a  shielded  equal  ratio-arm  bridge  which  is  satisfactory 
for  the  measurement  of  capacitance  by 
direct  comparison  with  a  standard. 
This  bridge  will  measure  mutual, 
grounded,  and  direct  capacitance.  The 
requirement  that  must  be  met  in  order 
to  measure  mutual  capacitance  is  that 
the  bridge  corners  C  and  D  be  balanced 
to  ground.  Since  there  is  no  capacitance 
to  ground  from  B,  and  since  A  and  C 
are  at  the  same  potential  when  the 
bridge  is  balanced,  this  requirement  is 
met  when  the  total  capacitance  from  A 
and  C  to  ground  equals  the  capacitance 
from  D  to  ground.  If  at  the  same  time 
these  capacitances  are  proportioned  to 
keep  the  bridge  in  balance,  then  the  ca- 
pacitances from  A  and  C  to  ground  will 
be  equal,  each  being  half  the  capacitance 
from  D  to  ground.  This  adjustment 
may  be  made  by  means  of  auxiliary  air 
capacitors.  Grounded  measurements 
are  made  by  grounding  the  D  corner  by 
switch  Ki.  If  the  bridge  is  limited  to 
grounded  measurements,  D  may  be  per- 

manently grounded,  and  the  double  shielding  of  the  transformer 
is  unnecessary. 

Conductance.  The  loss  in  a  capacitor  may  be  measured  as  a  series  resistance  or  as  a 
shunt  conductance.  Using  the  bridge  of  Fig.  8  a  standard  resistance  may  be  connected 
either  in  series  or  in  parallel  with  the  standard  capacitor  to  balance  this  loss.  For  wide 
ranges  of  capacitance  and  power  factor,  however,  the  range  of  resistance  required  in  either 
case  is  objectionable.  The  series  method  may  require  excessively  small  resistances,  and 
the  shunt  method  may  require  excessively  high  resistances.  A  modification  which  permits 
of  a  practical  resistance  range  is  to  shunt  both  arms  by  a  resistance  of  fairly  high  value 
and  to  reduce  the  resistance  across  the  standard  to  balance  any  loss  in  the  capacitor  under 
test.  Satisfactory  values  are  10,000  ohms  for  the  fixed  resistance  across  CD  and  a  variable 
resistance  range  from  0.01  ohm  to  10,000  ohms  across  AD.  This  allows  the  measurement 
of  conductances  from  0.0001  micromho  up.  If  TO  is  the  reading  of  r*n  in  ohms  for  the  zero 
balance  and  ri  is  the  reading  with  the  capacitor  in  circuit,  then 


FIG.  8.     Completely  Shielded  Equal  Ratio-arm  Bridge 
Suitable  for  Balaneed-to-groimd  or  Grounded  Measure- 
ments 


and  the  ratio  arms 


gx 


mhos 


If  TO  is  approximately  10,000  ohms,  and  the  conductance  is  below  1  micromho  T  this 
reduces  to 


gx 


.  , 

microrahos 


The  bridge  then  becomes  practically  direct  reading  for  conductance.  This  approxima- 
tion holds  for  conductance  values  encountered  in  all  good  standards  at  moderate  frequen- 
cies. These  expressions  are  actually  the  difference  in  the  loss  between  the  standard  and 
the  capacitor  under  test.  If  the  standard  has  conductance,  it  must  be  added  to  the 
difference  obtained. 

In  order  to  calibrate  standards  for  conductance,  it  is  necessary  to  have  a  primary  stand- 
ard of  zero  or  known  conductance.  This  is  usually  obtained  by  means  of  an  air  capacitor 
specially  designed  to  give  a  direct  capacitance  having  no  loss.  If  an  air  capacitor  is  built 


11-26 


ELECTKICAL  MEASUREMENTS 


so  that  each  set  of  plates  is  mounted  by  insulating  supports  on  a  third  metal  conductor 
or  shield,  we  then  have  a  system  of  three  capacitors  having  three  direct  capacitances  such 
as  shown  in  Fig.  7,  in  which  the  dielectric  loss  is  wholly  in  the  direct  capacitances  from 
the  plates  to  the  support,  the  direct  capacitance  between  the  plates  including  no  dielectric 
loss.  This  method,  or  the  use  of  a  variable  air  capacitor  with  conductance  independent 
of  setting,  is  the  common  method  for  obtaining  standards  of  zero  conductance. 

Accuracy.  The  accuracy  of  a  bridge  of  this  type  depends  on  the  accuracy  with  which 
the  capacitance  standards  are  known.  An  overall  accuracy  of  0.1  to  0.25  per  cent  for 
capacitance  is  possible  for  frequencies  up  to  1  megacycle.  The  conductance  accuracy 
stated  in  terms  of  power  factor  is  in  the  order  of  ±(1  per  cent  +  0.0001). 

DIRECT  CAPACITANCE.  Direct  capacitance  may  also  be  measured  on  the  bridge  of 
Fig.  8  as  follows:  Ground  the  point  C  of  the  bridge  and  all  conductors  of  the  apparatus 
under  test  except  the  two  terminals  between  which  the  direct  capacitance  is  desired. 
Connect  one  of  these  to  C  and  one  to  D,  and  obtain  a  bridge  balance.  Then  transfer  the 
terminal  connected  at  C  to  A,  and  rebalance  the  bridge.  A  little  consideration  will  show 
that  the  only  capacitance  transferred  in  this  operation  is  the  direct  capacitance  required, 
and  the  difference  between  the  .balances  is  therefore  equal  to 
~  twice  this  capacitance. 

A  second  method  which  is  satisfactory  where  all  the  capaci- 
tances involved  are  small,  such  as  the  interelectrode  capaci- 
tances of  a  vacuum  tube,  is  to  connect  the  capacitance  to  be 
evaluated  across  CD,  all  other  terminals  including  ground 
being  connected  to  B.  The  capacitance  thus  placed  across 
BC  will  affect  the  accuracy  of  conductance  but  not  of  capaci- 
tance measurement,  and  the  direct  capacitance  may  be 
measured  directly, 

The  modified  circuit  shown  in  Fig.  9  has  two  advantages 
for  the  measurement  of  direct  capacitance.  By  using  a  pair 
of  closely  coupled  inductors  for  ratio  arms,  that  is,  parallel 
windings  on  a  common  core,  the  effect  of  the  capacitance 
across  BC  will  be  very  much  reduced.  If,  in  addition,  the 
capacitance  standard  is  such  that  capacitance  is  transferred 

from  CD  to  J.D,  keeping  the  total  value  CA  +  Cc  constant,  and  a  fixed  capacitance  CD 
is  connected  in  series  in  the  D  lead,  it  may  be  shown  that  the  bridge  unbalance  correspond- 

ing to  an  unbalance  CA  —  Cc  is  equal  to       A       —          .    If  CD  is  made  small  compared 

- 


FIG.   9.     Bridge   for   Measure- 
ment of  Direct  Capacitance 


T  LC  T- 
Cc  +  CD,  very  small  values  of  direct  capacitance,  as  low  as  0.0001  j 


,  may  be 


with  CA 
measured. 

A  third  method  is  by  means  of  the  Wagner  ground. 

The  Wagner  Ground.  A  common  modification  of  the  simple  equal  ratio-arm  bridge 
and  of  other  bridges  involves  the  Wagner  ground.  This  is  a  simple  method  of  eliminating 
the  effect  of  stray  admittances,  compared  with  complete  shielding  of  the  bridge.  Referring 
to  Fig.  8,  but  with  power  source  and  detector  interchanged,  it  consists  of  a  potentiometer 
connected  across  the  input  corners  AC  of  the  bridge,  the  adjustable  contact  being  grounded 
and  adjusted  to  bring  the  D  corner  of  the  bridge  to  ground  potential.  In  some  bridges, 
complex  impedances  are  required  instead  of  resistances  for 
this  balance.  It  has  two  disadvantages.  First,  it  requires  for 
any  measurement  two  balances,  namely,  the  adjustment  of 
the  Wagner  ground  and  of  the  bridge  itself,  and  these  bal- 
ances are  not  independent.  This  makes  the  balance  pro- 
cedure relatively  slow  and  complicated.  Second,  the  quan- 
tity measured  is  the  direct  capacitance.  When  the  mutual  or 
grounded  capacitance  is  desired,  it  must  be  computed  from 
the  measured  values  of  the  individual  direct  capacitances. 
These  limitations  considerably  restrict  its  usefulness. 

THE  SCHERING  BRIDGE.  At  high  frequencies,  adjust- 
able resistance  standards  have  serious  limitations.  Above 
about  1  megacycle,  the  Schering  bridge  is  used  to  avoid  them. 
In  this  bridge  a  fixed  capacitance  and  a  fixed  resistance  are  FlG-  10- 
in  diagonally  opposite  arms,  and  the  other  arms  may  be 
arranged  to  allow  both  capacitance  and  conductance  to  be 
balanced  by  variable  capacitors.  The  circuit  and  balance  equations  are  given  in  Fig.  10. 
K  the  bridge  is  balanced  and  then  the  unknown  is  connected  across  C\  and  the  bridge 
rebalanced  by  means  of  Ci  and  C*,  the  change  in  Ci  equals  the  capacitance  of  the  unknown 
ftttd  tfae  change  in  Cs  is  inversely  proportional  to  its  conductance.  If  series  components 
are  desired  the  unknown  may  be  connected  either  in  series  or  in  parallel  with  Cs-  Then  Cg 


Schering  Bridge  for 
Measurement  of  Capacitive  Im- 
pedances 


INDUCTANCE  AND  EFFECTIVE  RESISTANCE          11-27 

balances  for  capacitance  and  Ci  for  series  resistance  but  the  bridge  will  not  be  direct  reading 
for  both  components. 

OTHER  METHOBS.  Bridge  methods  have  the  advantage  of  a  null  balance.  They  do 
not  have  the  advantage  of  both  grounded  input  and  grounded  output  unless  transformers 
are  used.  Other  methods  are  available,  particularly  at  high  frequencies,  which  have  both 
these  advantages.  They  consist  essentially  of  two  unbalanced  transmission  networks 
connected  in  parallel,  the  required  relation  between  the  arms  for  zero  output  being  that 
their  transfer  constants  be  equal  in  magnitude  but  180°  out  of  phase  (see  Section  6). 
Commercial  measuring  circuits  of  this  type  are  available,  known  as  shunted  T  and  twin/T 
circuits. 

The  most  popular  method  other  than  null  methods  is  the  simple  tuned  circuit,  the  con- 
dition sought  being  a  maximum  or  minimum  of  either  voltage  or  current  with  variation  of 
inductance,  capacitance,  or  frequency.  A  commercial  circuit  using  this  method  known 
as  a  Q  meter  is  described  in  article  13. 

Practically  all  these  special  circuits  substitute  a  capacitance  standard  for  the  unknown 
in  measuring  capacitance,  the  difference  between  them  lying  in  the  method  of  measuring 
the  loss.  Capacitance  is  seldom  measured  in  terms  of  other  quantities  owing  to  the 
superiority  of  capacitance  standards  over  all  others. 

13.  MEASTJEEMENT  OF  INDUCTANCE  AND  EFFECTIVE  RESISTANCE 

As  for  capacitance,  the  simplest  type  of  measurement  is  direct  comparison,  and  the 
bridge  of  Fig.  8  is  capable  of  the  same  accuracy  for  inductance  measurements  as  for  capaci- 
tance measurements,  limited  only  by  the  standards.  However,  inductance  standards  are 
inherently  less  satisfactory  than  capacitance  or  resistance  standards,  particularly  for  wide 
inductance  and  frequency  ranges.  Accordingly,  other  less  symmetrical  bridges  are  common 
in  order  to  take  advantage  of  standards  of  capacitance  and  resistance.  Of  these  the 
bridges  most  used  are  the  resonance  method  with  the  equal  ratio-arm  bridge,  the  Owen 
bridge,  the  Maxwell  bridge,  and  the  Schering  bridge. 

COMPARISON  METHOD.  This  measurement  is  made  by  comparing  an  inductance 
in  the  CD  arm  with  an  adjustable  standard  in  the  AD  arm  of  an  equal  ratio-arm  bridge 
such  as  shown  in  Fig.  8.  In  order  to  balance  the  effective  resistance  component  and  to 
measure  it  when  required,  it  is  necessary  to  add  a  series  resistance  in  either  the  CD  arm 
or  the  AD  arm,  depending  on  the  relative  resistance  values  of  the  unknown  impedance 
and  of  the  standard.  The  usual  procedure  is  to  connect  a  standard  resistance  by  means  of  a 
switch  at  D  to  throw  the  resistance  into  either  arm  as  required.  The  bridge  permits  bal- 
anced to  ground  and  grounded  measurements,  as  in  capacitance  measurements.  When 
adjustable  standard  inductances  are  used,  and  when  the  series  resistance  required  is  large, 
the  shielding  of  the  standards  becomes  cumbersome  and  the  bridge  is  commonly  used 
grounded. 

The  inductance  of  the  unknown  is  equal  to  the  inductance  of  the  standard  at  balance. 
The  effective  resistance  is  given  by  rx  =  TL  ±  rs  where  rj,  is  the  effective  resistance  of  the 
standard  inductance  and  r8  is  the  setting  of  the  standard  resistance,  the  sign  depending  on 
the  position  of  the  switch. 

The  accuracy  of  such  a  circuit  depends  on  the  frequency  and  on  the  accuracy  of  calibra- 
tion of  the  inductance  standards.  An  accuracy  of  about  0.25  per  cent  for  inductance  and 
2  per  cent  for  resistance  is  possible  for  frequencies  as  high  as  a  megacycle. 

Wagner  Ground.  This  can  be  used  with  the  above  method  to  avoid  shielding.  However, 
the  limitations  of  this  method  as  outlined  in  article  12  apply  equally  to  inductance  measure- 
ments. 

RESONANCE  METHOD.  Inductance  can  be  compared  with  capacitance  and  fre- 
quency either  by  series  or  parallel  resonance.  The  series  method  has  the  advantage  of 
giving  directly  the  series  resistance  and  inductance  of  the  coil  which  are  the  values  usually 
desired.  It  is  also  inherently  a  low-impedance  circuit,  and  this  is  often  an  advantage 
where  the  voltage  available  from  the  power  source  is  limited.  The  parallel  method  gives 
the  equivalent  parallel  values,  which  usually  require  subsequent  transformation.  It  is 
inherently  a  high-impedance  circuit. 

The  principle  on  which  resonance  measurements  are  based  is  the  adjustment  of  the 
capacitor  until  the  tuned  circuit  has  zero  or  infinite  reactance;  that  is,  it  is  equivalent  to  a 
pure  resistance.  The  measurement  is  usually  made  on  an  equal  ratio-arm  bridge,  but  any 
bridge  that  will  determine  when  the  impedance  of  the  circuit  is  a  pure  resistance  and  that 
will  measure  the  resistance  is  suitable.  The  principal  objections  to  the  method  are  that 
it  is  not  direct  reading  and  the  accuracy  is  dependent  on  the  frequency  of  the  source. 

Series  Resonance.  For  the  series  measurement  by  the  equal  ratio-arm  bridge  of  Fig.  8, 
an  adjustable  resistance  is  connected  in  the  AD  arm  and  the  capacitance  in  series  with  the 


11-28 


ELECTRICAL  MEASUREMENTS 


unknown  in  the  CD  arm.  Since  the  unknown  forms  only  part  of  the  impedance  in  the  CD 
arm,  balanced-to-ground  measurements  are  impractical  and  this  measurement  is  usually 
made  with  D  grounded.  The  capacitance  is  connected  from  C  to  the  unknown  in  order 
that  one  terminal  of  the  unknown  may  be  connected  to  ground.  The  balance  is  obtained 
by  adjusting  the  standard  resistance  and  capacitance,  and  at  balance  the  following  relation 
holds: 

oi2LzC4  =  1     or    Lx  —  -^7 
orCs 

rx  is  equal  to  r,  less  the  equivalent  series  resistance  of  C8.  If  the  conductance  of  C,  is 
used,  as  it  generally  is,  since  when  multiple  standards  are  used  their  conductances  add 
directly,  then 


- 

*      c, 

where  rx,  Lx  are  the  values  of  the  unknown. 

ga  is  the  conductance  of  the  standard  Ca. 

Parallel  Resonance.  For  this  measurement  the  capacitance  and  the  unknown  induct- 
ance are  connected  in  parallel  in  the  CD  arm  of  the  bridge.  Either  balanced-to-ground 
or  grounded  measurement  may  be  made.  Since  the  resistance  required  for  the  balance 
may  be  very  high,  an  arrangement  similar  to  that  for  a  capacitance  bridge  is  customary; 
that  is,  a  fixed  resistance,  usually  10,000  ohms,  is  connected  across  CD  and  the  loss  is  read 
as  conductance  by  means  of  a  variable  resistance  across  the  AD  arm  of  the  bridge.  The 
balance  is  then  obtained  in  the  same  way  as  for  series  resonance.  Then 

_  1  ,  ro  —  TI 


where  g»  is  the  conductance  of  the  condenser  C,, 

ro  and  ri  are  the  open-circuit  and  final  readings  of  TV 
Lp  is  the  parallel  inductance  of  the  unknown  impedance. 
gp  is  its  conductance. 

The  series  equivalents  may  be  obtained  by  transformation, 

Accuracy.    The  accuracy  of  resonance  bridges  depends  on  the  accuracy  of  both  the 
capacitance  standards  and  the  frequency.    Accuracies  as  high  as  0.1  per  cent  are  possible 

without  extreme  precautions.  They 
are  probably  the  most  accurate  cir- 
cuits for  measuring  effective  resist- 
ance, accuracies  of  the  order  of  2  per 
cent  being  readily  obtainable,  even 
for  high-Q  coils. 

THE  OWEN  BRIDGE.  This 
bridge  in  common  with  the  Max- 
well bridge  has  the  advantage  that 
inductance  is  measured  in  terms 
of  capacitance  and  resistance.  It 
is  not  frequency  sensitive  and  can 
be  made  direct  reading  for  both 
L  and  r.  A  shielded  circuit  for 
the  Owen  bridge  is  shown  in  Fig. 
11.  The  equations  of  this  circuit  at 
balance  are 


L  = 


and 


FIG.  11.     Shielded  Owen  Bridge  for  Measurement  of  In- 
ductance and  Effective  Resistance 

proportional  to  ; 


where  r2  includes  the  effective  resist- 
ance of  the  unknown.  Two  methods 
of  operation  are  possible.  Both  C\ 
and  n  may  be  adjustable,  and  then 
the  inductance  of  the  unknown  is 


and  the  total  effective  resistance  in  the  CD  arm  is  proportional  to  1/Ci. 
It  is  usually  more  convenient  to  have  Ci  fixed  or  adjustable  in  a  few  steps  and  to  have 
TZ  adjustable.  Then  by  taking  a  short-circuit  reading  r0  for  r2  it  follows  that  L  —  Cartf\ 
and  rx  =  ro  —  r$.  Since  the  unknown  is  not  connected  directly  across  the  arm  CZ>,  bal- 
anced-to-ground measurements  are  impractical.  Since  the  resistances  are  capable  of 
adjustment  to  their  nominal  values  with  a  high  degree  of  precision,  the  bridge  is  direct 
reading  for  both  inductance  and  resistance  without  the  need  for  calibration.  Also,  the 


INDUCTANCE  AND  EFFECTIVE  RESISTANCE          11-29 

range  of  inductance  may  be  made  very  wide  since  n  may  be  designed  to  have  as  many  as 
six  or  seven  dials.  The  bridge  is  not  so  statable  for  the  measurement  of  effective  resistance 
on  account  of  residual  reactances  in  the  various  bridge  arms  and  difficulties  in  shielding 
always  encountered  in  arms  having  series  impedances. 

The  shielding  is  made  only  partially  complete  in  the  interest  of  simplicity.  The  resist- 
ance TCD  is  used  to  compensate  the  shield  capacitance  across  JJ>,  a  resistance  being 
required  because  of  the  90°  phase  relation  of  the  ratio  arms.  With  this  circuit  inductance 
accuracies  as  high  as  0.1  per  cent  are  usual  at  audio  frequencies.  It  is  not  as  satisfactory 
at  higher  frequencies  as  the  following  bridges. 

THE  MAXWELL  BRIDGE.  In  this  bridge  fixed  resistors  are  used  in  diagonally  oppo- 
site arms.  The  inductance  and  effective  series  resistance  of  the  unknown  are  then  balanced 
by  capacitance  and  conductance  respectively  in  the  standard  arm.  The  circuit  and  balance 
equations  are  shown  in  Fig.  12.  If  ra  and  r&  are  not  pure 
resistances,  the  balance  equations  still  hold  if  their  phase 
angles  are  equal  but  of  opposite  sign.  If  gs  is  designed  to  read 
directly  in  conductance  (see  article  8)  the  bridge  may  be  made 
direct  reading  for  both  L  and  r.  This  feature  makes  the 
bridge  as  attractive  as  the  Owen  bridge.  In  addition,  since 
it  has  no  series  connections  in  the  arms,  stray  admittances 
may  be  compensated  more  readily  and  the  bridge  is  satis- 
factory at  higher  frequencies. 

THE  SCHERING  BRIDGE.  This  is  described  in 
article  12.  It  is  essentially  a  capacitance  bridge  but 
will  measure  inductance  as  a  negative  capacitance  and 
is  particularly  adapted  to  high-frequency  measurements  f10-  12-  Marwell  Bridge  for- 
of  low-Q  impedances  which  may  have  either  positive  or  M<Sd^th?aSrtSSM" 
negative  reactance. 

METERING.  The  values  of  inductance  and  resistance  of  magnetic-core  coils  are  usually 
a  function  of  the  saturation.  It  is  therefore  desirable  to  know  the  current  through,  or  the 
voltage  across,  the  coil  when  measured.  In  any  so-called  ratio-arm  bridge,  where  two 
adjacent  arms  are  invariable,  the  current  through  the  unknown,  or  the  voltage  across  it, 
will  have  a  definite  relation  to  the  total  bridge  current  or  voltage,  determined  only  by  the 
ratio  arms  and  the  choice  of  input  and  output  corners.  Thus  in  Fig.  8,  which  uses  a 
current  connection,  the  current  input  divides  at  balance  in  proportion  to  the  admittances 
of  the  ratio  arms,  and  a  meter  in  the  input  circuit  can  be  calibrated  to  read  directly  the 
current  through  the  unknown.  In  Fig.  11,  which  shows  a  voltage  connection,  tbe  input 
voltage  divides  at  balance  in  proportion  to  the  impedances  of  Ca  and  r&,  and  a  meter  across 
the  input  can  be  calibrated  to  read  directly  the  voltage  across  the  unknown.  Here  the 
calibration  will  be  a  function  of  frequency  but  is  independent  of  the  value  of  the  unknown. 

Bridges  of  the  so-called  product-arm  type,  having  the  fixed  impedances  diagonally 
opposite,  such  as  the  Maxwell  and  the  Schering  bridges,  do  not  have  this  feature,  and  the 
metering  is  usually  done  by  connecting  a  vacuum-tube  voltmeter  directly  across  the  un- 
known. 

SUPERIMPOSED  MEASUREMENTS.  Measurement  of  inductance  is  sometimes 
required  for  magnetic-core  coils,  with  direct  current  flowing  through  the  winding.  Such 
measurements  may  be  made  on  an  equal  ratio-arm  bridge  such  as  that  of  Fig.  S,  applying 
the  direct  current  across  the  BD  corners  and  separating  the  direct  current  from  the  alter- 
nating current  where  necessary  by  stopping  condensers  and  a  choke  coil.  A  more  con- 
venient bridge  is  the  Owen  bridge  of  Fig.  11.  If  direct  current  is  applied  across  BD  and 
alternating  current  across  AC,  the  only  additional  apparatus  required  is  a  stopping  con- 
denser in  the  detector  circuit.  The  Maxwell  bridge  is  also  well  adapted  to  making  these 
measurements. 

OTHER  METHODS.  At  very  high  frequencies  where  extreme  circuit  simplicity  Is 
desirable,  the  simple  tuned  circuit  is  commonly  used.  The  method  of  measurement  is  as 
follows. 

If  a  voltage  is  applied  to  an  inductance  in  series  with  a  capacitance  as  shown  in  Fig.  13 
and  either  L,  C,  or  co  is  varied  to  give  maximum  voltage  across  C,  then 

oji  =  — 

coC 
and 

Ez       <*>L          1 
Ei  ~~    r 

where  E\  is  the  voltage  across  the  tuned  circuit. 
&i  is  the  voltage  across  L  or  C. 
r  is  total  resistance  of  the  circuit. 


11-30  ELECTEICAL  MEASUREMENTS 

Thus,  if  the  values  of  EI,  E*,  and  w  are  measured,  the  inductance  and  resistance  of  a 
coil  can  be  determined  if  the  capacitance  and  resistance  of  the  capacitor  are  known,  and 
vice  versa.  A  refinement  of  this  circuit  consists  of  adjusting  the  input  voltage  to  a  definite 
value  either  directly  by  a  voltmeter  or  by  adjusting  the  current  into  the  known  resistance 
rc.  Then  the  voltmeter  across  G  may  be  calibrated  in  terms  of  Q.  If  the  capacitor  C 
has  negligible  loss,  this  will  be  the  Q  of  the  coil.  Commercial  measuring  sets  called  Q- 
meters  are  available  with  self-contained  oscillator,  low-loss  variable-capacitance  standard, 
and  vacuum-tube  voltmeter,  which  are  direct  reading  for  Q.  They  are  satisfactory  up 
to  200  megacycles  or  higher. 

There  are  a  number  of  errors  in  the  Q  reading  that  may  be  corrected  for  when  known. 
The  resistance  r  of  the  circuit  includes  the  equivalent  series  resistance  of  the  capacitor, 

which  includes  its  dielectric  loss  and  the  admit- 
tance of  F2-  The  frequency  response  of  the 
voltmeters  may  not  be  flat.  If  the  input 
current  is  controlled,  EI  is  affected  by  change 
in  impedance  of  rc  due  to  reactance  or  skin 
effect,  and  by  the  shunting  effect  of  r.  These 
errors  vary  independently  with  Q,  X,  and  &>  so 
much  that  no  average  figure  for  accuracy 
applies.  For  values  of  Q  about  100,  and  values 
FIG.  13.  Q  Meter  Circuit  of  X  about  100  ohms,  accuracies  of  5  per  cent 

can  be  expected  up  to  25  megacycles. 

MUTUAL  INDUCTANCE.  Mutual  inductance  between  two  coils  can  be  determined  by 
measuring  the  self-inductance  of  the  two  windings  by  any  suitable  method,  with  the  wind- 
ings connected  first  series  aiding  and  then  series  opposing.  The  difference  between  the 
two  values  is  four  times  the  mutual  inductance.  The  ground  conditions  are  usually  dif- 
ferent for  the  two  measurements  from  the  actual  operating  conditions.  This  may  be  a 
source  of  error,  particularly  where  the  coupling  is  low.  The  ratio  of  secondary  voltage  to 
primary  current  may  be  determined  directly  by  thermocouple  and  vacuum-tube  voltmeter. 

14.  SIGNAL  GENERATORS  AND  DETECTORS 

The  following  discussion  is  limited  to  the  special  requirements  which  apply  to  use  with 
measuring  circuits  such  as  those  described.  More  complete  information  may  be  found 
in  articles  36  to  40. 

SIGNAL  GENERATORS.  Many  single-frequency  generators  are  available  in  the 
audio-frequency  range  provided  the  requirements  are  not  severe.  However,  the  tuning 
fork  operated  by  a  microphone  or  vacuum  tube,  and  the  rotating  generator,  represent 
practically  the  only  types,  other  than  vacuum-tube  oscillators,  with  satisfactory  charac- 
teristics for  precise  work.  Where  a  range  of  frequencies  is  necessary  and  particularly  for 
frequencies  above  the  audio  range,  vacuum-tube  oscillators  are  used  almost  exclusively. 

VACUUM-TUBE  OSCILLATORS.  The  principal  requirements  for  an  oscillator  for 
measurement  purposes  are  adequate  output  level,  low  level  of  harmonics  and  other  spurious 
frequencies,  and  high  stability  of  frequency  and  output  level  with  respect  to  time  and 
temperature.  The  emphasis  on  these  requirements  depends  on  the  type  of  measurement. 
The  output  level  should  be  high  enough  to  insure  that  the  input  level  to  the  detector  is 
above  thermal  noise  for  the  most  precise  balance.  An  output  of  0.1  watt  is  usually  ade- 
quate. Level  stability  requirements  are  more  lenient  for  null  measurements.  Harmonic 
requirements  are  more  lenient  for  null  measurements  which  are  not  frequency  dependent. 
In  general,  when  untuned  detectors  are  used,  harmonics  should  be  held  below  3  per  cent 
of  the  fundamental,  and  for  null  resonant  measurements,  below  1  per  cent.  Harmonics 
may  be  suppressed  by  filtering  external  to  the  oscillator,  either  before  or  after  the  measuring 
circuit.  The  former  is  preferable  in  so  far  as  it  prevents  production  of  false  signal  by  modu- 
lation in  the  measuring  circuit,  but  it  places  severe  modulation  requirements  on  the  filter. 

Oscillators  are  in  general  of  three  types,  according  to  the  type  of  frequency-selecting 
network  used,  namely,  crystal,  LC,  and  rC  oscillators.  The  crystal  oscillator  is  the  most 
stable  and  is  the  preferred  type  for  fixed  frequency  applications  over  the  frequency  range 
for  which  crystals  are  applicable,  about  10  kc  to  10  me.  The  LC  oscillator  is  the  most 
versatile  and  can  be  used  over  the  whole  frequency  range  up  to  the  maximum  frequency 
at  which  lumped  constants  are  practical.  At  low  frequencies,  the  necessity  of  using  coils 
of  large  physical  size  to  obtain  a  high  Q  makes  it  cumbersome,  and  the  rC  oscillator  is 
preferred.  This  oscillator  has  the  advantages  that  the  components  are  small  even  at  very 
low  frequencies,  and  it  may  be  made  direct  reading  more  readily.  It  is  suitable  for  fre- 
qT*eiicies  lower  than  1  cycle  up  to  about  100  kc,  where,  in  spite  of  lower  stability,  it  com- 
petes mta  tiie  LC  oscillator  because  of  the  direct-reading  feature. 


BIBLIOGRAPHY  11-31 

HETERODYNE  OSCILLATORS.  These  oscillators,  having  an  output  frequency 
which  is  the  difference  between  the  frequencies  of  a  fixed  and  a  variable  oscillator,  can  be 
made  to  cover  a  wide  frequency  range  with  a  single  continuous  control.  This  has  two 
advantages:  it  allows  them  to  be  made  direct  reading  over  wider  ranges  than  the  conven- 
tional LC  oscillator,  and  the  broad  continuous  range  makes  them  suitable  for  sweep 
frequency  measurements  required  in  recording,  and  in  cathode-ray  visual  indication,  of 
frequency  characteristics. 

DETECTORS.  Detectors,  as  distinct  from  actual  measuring  instruments,  are  limited 
to  the  detection  of  null  balances  and  equality  of  output  from  different  circuits  or  from 
different  arrangements  of  the  same  circuit. 

The  principal  requirements  are  adequate  sensitivity  for  the  accuracy  required,  sufficient 
discrimination  against  harmonics,  and  stability  of  gain  with  time  and  temperature. 
Sensitivity  can  usually  be  obtained  by  amplification.  The  limit  is  determined  by  thermal 
noise,  which  depends  on  the  band  width.  This  limit  is  approximately  10~20  watt  per  cycle 
band  width.  For  instance,  using  a  detector  with  a  1000-cycle  band  width  working  out  of  a 
bridge  of  1000  ohms  impedance,  minimum  signal  to  exceed  thermal  noise  will  be  10 ~17 
watt  or  0.1  }j.voli. 

The  harmonic  suppression  required  depends  on  the  oscillator  harmonics,  the  method  of 
measurement,  and  the  characteristics  of  the  unknown.  Gain  stability  requirements  are 
least  severe  for  null  measurements.  They  are  most  severe  in  adjusting  two  outputs  to 
equality  using  some  form  of  suppression. 

The  telephone  receiver  with  or  without  preceding  amplification  is  the  simplest  and  most 
sensitive  detector  within  the  audio-frequency  range.  Owing  to  the  frequency  charac- 
teristic of  the  ear  and  of  the  receiver,  considerable  discrimination  against  harmonics  can 
be  obtained  in  the  frequency  range  where  they  are  most  sensitive.  At  higher  frequencies, 
the  heterodyne  type  of  detector  giving  an  audio  output  may  be  used  with  the  receiver. 
It  has  the  advantage  of  giving  considerable  discrimination.  In  all  detectors  care  must  be 
taken  that  the  harmonics  do  not  overload  the  input  sufficiently  to  cause  a.  false  signal  due 
to  modulation. 

In  addition  to  the  telephone  receiver,  a  rectifier,  such  as  copper  oxide  or  a  vacuum  tube, 
may  be  used  with  a  d-c  instrument  or  cathode-ray-tufoe  indicator.  These  require  greater 
amplification  and  more  discrimination. 

BIBLIOGRAPHY 

Campbell,  A.,  and  Childs,  E.  C-,  The  Measurement  of  Inductance,  Capacitance,  and  Frequency,  Macmil- 

lan  &  Co.,  London,  1935. 
Edgecumbe,  K.  W.  E.,  and  Ockenden,  F.  E.  J.,  Industrial  Electrical  Measuring  Instruments,  Pitman, 

London,  1933. 

Terman,  F.  E.,  Radio  Engineers  Handbook,  McGraw-Hill,  New  York,  1943. 
Chaffee,  E.  L.,  Theory  of  Thermionic  Vacuum  Tubes,  McGraw-Hill,  New  York,  1933. 
Moullin,  E.  B.,  Radio  Frequency  Measurements,  Griffin,  London,  1926. 
Hund,  A.,  High  Frequency  Measurements,  McGraw-Hill,  New  York,  1933. 
Hague,  B.T  Alternating  Current  Bridge  Methods,  Pitman,  London,  1923. 
Glazebrook,  Sir  R.,  Dictionary  of  Applied  Physics,  Macmillan  &  Co.,  London,  1923. 
Hartshorae,  L.,  Radio  Frequency  Measurements  by  Bridge  and  Resonance  Methods,  Chapman  and  Hall, 

London,  1941. 

Henney,  K.,  The  Radio  Engineering  Handbook,  McGraw-Hill,  New  York,  1941. 
Rider,  Jr.,  Vacuum  Tube  Voltmeters,  J.  F.  Rider,  New  York,  1941. 
Bur.  Standards  Circ.  74,  Radio  Instruments  and  Measurements. 
Grondahl,  L.  O.,  The  Copper-cuprous-oside  Rectifier  and  Photoelectric  Cell,  Rev.  Modern  Physics,  VoL 

5T  141  (1923),  including  extensive  bibliography  on  copper  oxide  rectifiers. 
Aiken,  C.  B.,  Theory  of  Diode  Voltmeters,  Proc.  I.R.E.,  Vol.  26,  859  (1938). 

Miller,  J.  H.,  Thermocouple  Ammeters  for  Ultra  High  Frequencies,  Proc.  I.R.E.,  Vol.  24,  1567  (1936). 
Behr,  L.,  and  Tarpley,  R.  E.,  Design  of  Resistors  for  High  Frequency  Measurements,  Proc.  I.RJE., 

Vol.  20,  1101  (1932). 

Campbell,  G.  A.,  The  Shielded  Balance,  Elec.  World,  Vol.  43,  647  (1904). 

Ferguson,  J.  G.,  Shielding  in  High  Frequency  Measurements,  Trans.  A.I.E.E.,  Vol.  48,  1286  (192§>. 
Campbell,  G.  A.,  Measurement  of  Direct  Capacities,  Bell  Sys.  Tech.  J".,  Vol.  1,  18  (1922). 
Christopher,  A.  J.,  and  Kater.  J.  A.,  Mica  Capacitors  for  Carrier  Telephone  Systems,  Trans.  AJ.E.E^ 

Vol.  65,670  (1946). 
Cone,  D.  I.,  Bridge  Methods  for  Alternating  Current  Measurements,  Trans.  A.I,E.E.t  Vol.  39,  1743 

(1920). 
Behr,  L.,  and  Williams,  A.  J.,  The  Campbell-Shackelton  Shielded  Ratio  Box,  Proc.  I.R.E.,  Vol.  20, 

969  (1932). 
Wagner,  K.  W.,  Zur  Messung  dielektrischer  Verluste  mit  der  Wechsektrombrucke,   Elekt.    Zeits., 

32  Jahrgang,  pp.  1001-1002. 

Kupfmuller,  Von  K-,  tlber  eiae  techmsche  Hochfrequenz  Messbmcke,  Elek.  Mack.  Tech.,  1925,  p.  263. 
Ferguson,  J.  G.,  Classification  of  Bridge  Methods,  Trans.  A.I.E.E.,  VoL  52,  861  (1934). 
Wilhelm,  H.  T.,  Impedance  Bridge  with  a  Billion  to  One  Range,  BeU  Td.  Rec.r  March  1945. 
Caial&gs  of  the  Leeds  and  Noorthrup  Co.,  General  Radio  Co.,  Westinghouse  Corp.,  aad  Weaton  Electrical 

Instrument  Co, 
General  Radio  Experimenter* 
Definitions  of  the  A.S.A. 


11-32 


ELECTRICAL  MEASUREMENTS 


WIRE  LINE  MEASUREMENT 

By  H.  J.  Fisher 


Attenuator 
FIG.  1.     Comparison  Method  for  Measuring  Insertion  Loss 


15.  TRANSMISSION  MEASUREMENTS 

Transmission  measurements  evaluate  a  circuit  or  facility  (wire  line,  cable,  radio  circuit, 
three-  or  four-terminal  network,  etc.)  in  regard  to  its  ability  to  transmit  telegraph,  voice^ 
modulated  carrier,  television,  or  other  communication  signals.  This  evaluation  usually 
involves  two  criteria,  (1)  effect  on  signal  amplitude  and  (2)  effect  on  signal  shape.  It  has 

been  found  convenient  to 
express  the  criteria  in 
terms  of  steady-state  mea- 
surements of  loss  or  gain, 
phase  shift,  and  envelope 
delay  distortion  vs.  fre- 
quency, and,  in  addition, 
non-linear  distortion 
(compression,  expansion, 
interehannel  modulation 
cross-talk)  as  a  function  of 
signal  amplitude.  Usu- 
ally the  device  or  circuit 
to  be  measured  is  one  of  many  connected  in  tandem  between  the  original  signal  source 
and  the  final  receiver,  and  it  is  desired  to  know  the  effect  on  overall  transmission  caused 
by  inserting  the  portion  in  question.  Since  the  individual  portions  of  a  system  are  usually 
designed  to  have  a  nominally  constant  impedance  (vs.  frequency)  of  a  standardized  value 
(e.g.,  600  «,  135  co,  75  w)  the  insertion  transmission  can  be  measured  directly  with  test 
equipment  having  the 
same  nominal  impedance. 
Impedance  variations 
with  frequency  of  the  unit 
to  be  measured,  of  the 
connected  circuits,  or  of 
the  testing  equipment, 
will  produce  errors  in  the 
insertion  transmission 
measurement  which  must 
be  considered.  If  all  im- 
pedances are  designed  to 
have  less  than  5  per  cent 
reflection  coefficient  these 


Avc  maintains  constant  voltage  wh.ic.tL  fs 
equivalent  to  zero  impedance  generator 


errors  usually  can  be  ne- 
glected. 

INSERTION  LOSS  OR  GAIN. 


o 

ra 

t                            System  or  um 

t  to  be  measured 

| 

r< 

5i>i    z 

^*"^*2 

~1> 

S~ 

N 

D-bin 
mfiter 

1 

y          »"*" 

I 

1 

f 

V 

^ 

H   Avc 

r~  " 
Ifc 

*  * 

vlW  (0-dbm) 
reference 
standard 

^n 

IMW  (0-dbm) 
reference 
standard 

Ibrating 
needed 

Sending  end  calibrating 
circuit  used  as  needed 

1  IMW  (Q-dbrn)| 
[sending  source) 

Receiving  end  caf 
circuit  used  as 

FIG.  2A.    Straightaway  Method  for  Measuring  Insertion  Gain  or  Loss 


Comparison  Method.  Figure  1  shows  a  commonly 
used  setup  for  measuring  insertion  loss.  The  attenuator  is  adjusted  until  equal  readings 
are  obtained  on  the  meter  for  the  two  positions  of  the  key.  When  measuring  gain  the 
attenuator  is  connected  in  tandem  with  the  unknown.  Systems  or  units  having  input 
frequencies  that  differ  from  the  output  frequencies  such  as  modulators  or  mixers  can  be 

measured  with  this  setup  provided  that  the 
detector  and  attenuator  characteristics  do  not 
change  over  the  frequency  range  of  interest. 
Straightaway  Method.  Figure  2A  shows  a 
setup  for  making  this  type  of  measurement. 
A  circuit  of  this  type  is  required  when  the  in- 
put and  output  terminals  are  not  available  at 
the  same  location.  The  circuit  is  self-explan- 
atory except  for  the  following  items. 

Dbm  Meter.    This  is  usuallv  a  broad-band 


FIG.  2B. 


Rectifiers 
DBM  Meter 


amplifier  having  adjustable  gain  in  10-db  steps  followed  by  a  linear  rectifier  (diode  or 
varistor)  with  a  meter  calibrated  in  dbm  over  about  a  13-db  range.    See  Fig.  2JB. 

1  Milliwatt  (0  Dbm}  Reference  Standard.  This  is  usually  a  thermocouple  circuit  of  the 
correct  impedance  arranged  to  be  calibrated  by  standard  d-c  power  as  determined  by  a  d-c 
milliammeter. 


TRANSMISSION  MEASUREMENTS 


11-33 


Voltage  Method.  For  making  measurements  on  working  systems  without  causing  dis- 
turbance to  the  working  signals,  a  frequency  selective  voltmeter  is  used  (see  reference  3), 
This  is  a  high-impedance  selective  heterodyne  detector  of  adjustable  sensitivity  in,  say, 
10-db  steps  followed  by  a  linear  detector  and  dbm  meter.  The  measurements  are  usually 
made  at  the  pilot  frequencies  or  other  frequencies  not  in  the  signal  channels.  Since  the 
working  signals  may  be  at  higher  levels  than  the  pilots,  it  is  important  that  thej&rst 
modulator  and  input  amplifier  be  operated  at  levels  low  enough  to  reduce  the  error  due  to 
inter  modulation  products  falling  at  the  measuring  frequency.  Frequency  discrimination, 
at  intermediate  and  final  frequency  stages,  sufficient  to  eliminate  all  other  unwanted 
frequencies  is  obtained  by  means  of  quartz  crystal  filters  and  selective  interstage  circuits. 
The  required  discrimination  can  be  reduced  by  about  26  db  by  the  use  of  a  linear  rectifier 
rather  than  a  peak  detector  and  about  23  db  with  square-law  or  thermal-type  indicators. 

MEASUREMENT  OF  INSERTION  TRANSMISSION  (LOSS  OR  GAIN)  OF  CIR- 
CUITS AND  UNITS  HAVING  MISMATCHED  IMPEDANCES.  As  stated  above, 
when  the  reflection  coefficient  of  the  circuit  or  unit  being  measured  and  the  test  equipment 
are  less  than  5  per  cent  when  re- 


rJH 


Line  amplifiers 


453  kc  (A) 

-57  kc  (B) 


lOOO'v 

loob'v 


FIG.  3 A.     Modulation   Measurements    on 
Telephone  System 


ferred  to  the  nominal  impedance 
the  measured  insertion  transmis- 
sion is  usually  negligibly  different 
from  the  true  insertion  transmis- 
sion. In  some  broad-band  systems 
the  impedances  of  the  tandem, 
components  vary  considerably 
with  frequency  from  the  nominal, 
and  the  reflection  losses  incurred 
are  sometimes  employed  in  overall 
equalization  of  the  system.  When 
measuring  components  of  such  a 
system  with  nominal  impedance 
test  equipment  there  is  a  discrep- 
ancy between  the  measured  and 
actual  insertion  transmission.  When 
measuring  with  a  voltmeter  type 
of  circuit  the  discrepancy  is  differ- 
ent and  usually  larger.  This  in  itself  causes  no  great  complication  in  the  maintenance 
of  these  systems  since  the  operating  limits  are  specified  having  in  mind  the  type  of  test 
equipment  which  is  used.  Difficulty  does  arise  when  attempts  are  made  to  correlate 
measurements  made  by  the  two  methods.  A  calculation  using  complete  impedance  in- 
formation and  rigorous  insertion  transmission  equations  is  necessary. 

INTERMODTJLATION  DISTORTION  MEASUREMENT.     As  a  result  of  non-linear 
distortion  (such  as  occurs  in  vacuum-tube  amplifiers)  in  a  transmission  system,  frequen- 
cies other  than  those  applied  to  the 
I  Line  section  j  — *49  kc  (2  A-B) 


10QO<v 


DBM  meter 
Overall    Carrier 


-»-49  kc  (2  A-B) 
-*~53  kc  (A) 
-»-57  kc  <B) 


including     . 

amplifiers    j  ~ **53  kc  (A) 


Sectionalizing  filter 
suppresses  49  kc 


7 


input  of  the  system  are  produced.  For 
example,  if  two  tones  are  applied  to 
the  system,  say  (a)  and  (6),  new  com- 
ponents having  frequencies  such  as  2a, 
26,  a  =t  6,  2a  ±  6,  26  ±  a,  3a,  36.,  etc. 
are  produced.  In  some  cases,  these 
new  frequencies  fall  outside  the  band 
of  interest  and  need  not  be  considered, 
but  in  wide-band  systems  many  of 
these  products  fall  within  the  trans- 
mitted band.  In  multichannel  sys- 
tems these  new  frequencies  may  result 
in  interchannel  interference.  To  pre- 
vent this  interference  from  exceeding 
allowable  amounts,  measurement  of  intermodulation  distortion  is  a  necessary  part  of  the 
maintenance  of  multichannel  systems. 

There  are  two  types  of  test:  (1)  an  over-all  test  to  check  whether  the  circuit  as  a  whole 
meets  specified  requirements;  (2)  a  test  on  a  portion  of  the  circuit  or  on  individual  re- 
peaters to  locate  the  defective  tubes  or  other  component  or  a  maladjustment. 

Figure  3A  shows  how  the  over-all  test  is  made  on  a  typical  12-channel  carrier  system. 
Channels  1  to  9  are  continued  in  operation,  channels  10,  11,  and  12  being  turned  down. 
Channels  11  and  12  are  energized  with  1000-cycle  power  of  specified  level,  and  after  pass- 
log  through  the  terminal  modulating  equipment  they  appear  on  the  line  as  53  and  57  kc. 


Selective  detector 
including  DBM  meter 

Preselection  'filter 
passes  49  kc 

PIG.  3B.     Modulation  Measurements  on  Line  Section 


11-34 


ELECTRICAL  MEASUREMENTS 


As  a  result  of  third-order  intermodulation  in  succeeding  amplifiers,  49  kc  (2a  —  o)  is  pro- 
duced. At  the  receiving  end  this  product  appears  in  voice  channel  10  as  1000  cycles, 
where  it  is  measured  with  a  sensitive  dbm  meter.  The  2a  —  b  product  is  used  for  this  test 
because  it  is  the  product  most  likely  to  be  excessive.  The  reason  for  this  is  that  the  modu- 
lation component  produced  at  each  repeater  tends  to  add  in  phase  with  the  components 
produced  by  other  repeaters  whereas  most  other  products  add  on  a  random  basis. 

To  localize  sources  of  excessive  intermodulation  the  line  may  be  sectionalized  as  shown 
in  Fig.  3B.  In  this  case,  the  three  channels  are  turned  down  as  before,  the  53  and  57  kc 
originating  as  1000  cycles  at  the  transmitting  terminal.  Any  49  kc  produced  between  the 
terminal  and  the  suppression  filter  is  suppressed  and  the  test  is  essentially  originated  at 
that  point.  The  selective  detector  and  preselection  filter  are  portable  and  can  be  moved 
as  near  to  the  suppression  filter  as  desired,  for  example  close  enough  to  include  only 
one  repeater- 

In  other  multichannel  systems  the  modulating  test  tones  are  applied  directly  to  working 
lines  by  means  of  high-frequency  oscillators.  They  are  allocated  in  spaces  between  work- 
ing channels  and  are  also  chosen  so  that  the  product  being  measured,  which  may  be 
2a,  3a,  2a  —  6,  etc.,  also  falls  in  an  idle  part  of  the  spectrum. 

See  references  1  to  3. 

INSERTION  PHASE  MEASUREMENT.  This  measurement  is  almost  entirely  re- 
stricted to  the  laboratory,  where  it  is  particularly  useful  in  connection  with  the  measure- 
ment of  the  feedback  factor  (/*$)  loop  of  a  feedback  amplifier  (see  references  4  and  5)  and 


Circuit     >impedance    BO-/  Differentia! 

ndertestl  (        probe  '     7 v  <  voltmeter 


Rectifie 


Differentia!  meter 

response= 

K  (rectified  sum- 

rectiiied  difference). 


FIG.  4J..     Direct-reading  Phase-measuring  Circuit 

the  insertion-  phase  of  networks  (see  reference  6).  In  connection  with  transmission  of 
television  signals  over  coaxial  cables  phase  data  are  obtained  indirectly  by  the  integration 
of  the  envelope  delay  characteristic. 

Figure  4J.  shows  a  direct-indicating  type  of  circuit  by  means  of  which  the  ju$  character- 
istic can  be  quickly  obtained.  Attenuators  A  and  B  are  adjusted  so  that  the  input  and 
output  vectors  are  equal  in  magnitude.  The  loss  or  gain  of  the  circuit  under  test  can  be 
obtained  from  these  attenuator  readings.  By  an  alternative  arrangement,  using  avc 
amplifiers  this  can  be  accomplished  automatically.  Assuming  that  vectors  of  equal  mag- 
nitude are  applied  to  the  hybrid  coil  the  meter  reading  will  indicate  the  phase  difference 
directly.  Figure  4B  shows  the  characteristics  of  the  direct-indicating  phase  indicator. 
The  difference  of  the  rectified  sum  and  difference  outputs  is  used  because  it  permits  the 

r/7T          CX      ~ 
response  =  kf  cos  (  -  -f-  - 


By 

tltis  mean®  a  quite  linear  indication  of  phase  difference  can  be  obtained  over  a  ISO0  range 
and  wiikin  the  range  of  90°  ±  45°  the  deviation  is  usually  negligible.    This  range  of  good 


TRANSMISSION  MEASUREMENTS 


11-35 


-H.O 


0  40          80         120       ISO 

dboz=angJe  between  vectors,  degrees 

FIG.  4B.     D-c  Response  vs.  Differ- 

ence  in  Phase  of  Two  Equal-frequency 

Sine  Waves 


linearity  may  be  shifted  to  any  other  desired  portion  of  the  range  by  means  of  a  calibrated 

phase  shifter  in  one  of  the  branches.     In  Fig.  4A  a 

modulation  method  is  shown  to  provide  selectivity  to 

reduce  the  effect  of  noise  and  to  permit  the  use  of 

fixed-frequency  circuits,  but  successful  broad-band  sets 

have  also  been  constructed.     This  type  of  set  is  also 

easily  adapted  to  the  automatic  recording  of  ju£. 

Other  types  of  phase-measuring  circuits  have  been 

devised.    Generally  they  require  the  adjustment  of  the 

reference  and  unknown  vectors  to  equal  magnitude. 

Phase  may  then  be  computed  from  the  difference  in 

magnitude  of  their  sum  and  difference,  or,  by  means  of 

an  adjustable  calibrated  phase  shifter  (of  any  of  several 

well-known  types)  placed  in  either  branch,  the  sum  or 

difference  may  be  adjusted  to  a  null  and  then  the  phase 

shift  may  be  read  directly  from  the  calibrated  phase 

shifter,  due  allowance  being  made  for  quadrant  deter- 
mination.   In  a  variation  of  this  method  suggested  by 

S.  T.  Meyers  and  used  extensively,  the  sum  or  difference 

is  adjusted   by  means  of  a  calibrated  phase  shifter 

to  equality  with  the  reference  and  unknown  vector.    This  indicates  a  120°  phase  difference 

between  the  two  applied  vectors  which  when  added  to  or  subtracted  from  the  indicated 

phase  shift  depending  on  the 
quadrant  gives  the  actual 
phase  difference. 

Another  widely  used  phase- 
measuring  circuit  is  described 
in  W.  P.  Mason's  patent  U.  S. 
1,684,403.  This  is  known  as 
the  sum-and-d'iff erence 
method  and  requires  only 
commonly  available  equip- 
ment such  as  an  oscillator,  a 
detector,  and  attenuators  (see 
reference  6). 

INSERTION  ENVELOPE 
DELAY  DISTORTION 
MEASUREMENT.  In  some 
types  of  communication,  such 
as  television,  telephoto,  and 
telegraph,  distortion  of  signal 
shape  is  more  important  than 
for  ordinary  telephone  com- 


Ehvebpe  tkJay  distortion 
indicating  meter 


LuoearN 
amp.  reck 


Fia.  5.     Envelope  Delay  Distortion  Measuring  Circuit 


munication  (see  references  7—12).  Equally  as  important  as  attenuation  distortion  (vs. 
frequency)  as  a  criterion  for  distortion  of  signal  shape  is  envelope  delay  distortion.  Enve- 
lope delay  is  expressed  as  dfi/du,  and  in  an 
ideal  system  it  is  independent  of  frequency 
in  the  range  of  interest.  Envelope  delay  dis- 
tortion is  corrected  by  means  of  phase-shift 
networks,  the  objective  being  to  obtain  a 
phase  shift  vs.  frequency  characteristic  in  the 
frequency  range  of  interest  having  a  slope 
dfi/d<u  which  is  constant.  For  networks  and 
amplifiers  the  necessary  phase  data  are  usu- 
ally obtained  directly  from  phase  measure- 
ment. For  long  lines  it  is  difficult  to  make 
accurate  straightaway  phase  measurements 
on  account  of  the  instability  of  the  loss  and 
phase  of  the  line  which  may  include  many 
repeaters,  and  the  usual  practice  is  to  measure 
envelope  delay  distortion.  From  these  data 
the  phase  requirements  for  the  equalizer  can 
be  computed. 


Envelope  delay 

distortion,  ~~ 

jft\rTQ<zf*rnritfo 


irf 


FIG.  6. 


Typical  Phase  Characteristic  of  a  Trans- 
mission Line 


Figure  5  shows  a  circuit  for  making  straightaway  measurements  of  the  envelope  delay 
distortion  of  long  coaxial  circuits.    At  the  sending  end  four  frequencies  of  equal  amplitude 


11-36  ELECTRICAL  MEASTJBEMENTS 

are  transmitted:  /i  +  p/2  and  fi  —  p/2,  representing  the  reference  signal;  and  /2  +  p/% 
and  fz  —  p/2,  representing  the  test  signal.  Referring  to  Fig.  6,  assume  that  the  line  under 
test  has  a  phase  vs.  frequency  characteristic  as  shown.  At  the  distant  end  of  the  line  each 
of  the  four  frequencies  will  have  been  shifted  in  phase  by  different  amounts  j3(i_p/j), 
£d  +  p/s>»  ftt-p/2),  and  0(1+  p/2).  Now  we  can  say  that 

Aft   =   £(I+p/2)   -   |8(l_p/2),  A02  =   /3(2-f-j>/2)   —   /3(2_p/2> 

and  if  p  is  chosen  small  enough  then  for  all  practical  purposes 

Aft       dft  A&  __  d& 

_,  —  =  __  —    and    r       —  T 
2?rp        ao>  27rp       oco 

The  envelope  delay  distortion  f-p  -  —  *  J  is  equal  to  -    g6Q  microseconds. 


If  /2  is  varied  over  the  band  of  interest  the  measurement  of  A&  —  Aft  will  represent 
envelope  delay  distortion  referred  to  the  envelope  delay  at  the  reference  frequency  /i.  If 
the  set  is  given  a  "zero"  adjustment  by  means  of  the  "zero  adjust"  phase  shifter  so  that 
with  a  distortionless  line  or  resistance  pad  in  place  of  the  line  under  test  the  indicated 
envelope  delay  distortion  is  zero  then  the  equipment  at  the  receiving  end  will  provide  an 


indication  proportional  to  (Aft  —  Aft)  and  therefore  also  to  I —  )  ,  the  envelope 

\O6)        aw  / 

delay  distortion.  The  receiving  circuit  functions  as  follows:  the  reference  signal  and  the 
test  signal  are  separated  by  filters  as  shown  and  then  demodulated  to  obtain  the  difference 
products.  The  outputs  of  the  two  demodulators  are  of  the  same  frequency  p  but  have  a 
phase  difference  equal  to  Aft  —  Aftt  assuming  that  the  zero  adjustment  has  been  made. 
The  amplitudes  of  the  two  vectors  are  made  equal  by  means  of  the  attenuators  (which 
incidentally  provides  a  measure  of  attenuation  distortion)  and  combined  in  the  hybrid 
coil  from  which  two  new  vectors  are  obtained,  one  whose  amplitude  is  a  function  of  the 
sum  of  the  two  vectors  (^Afe  +  l?4/Ji)  and  the  other  whose  amplitude  is  a  function  of  the 
difference  (EAfo  —  EA&I)  .  These  new  vectors  are  rectified  separately  in  linear  amplifier 
rectifiers,  and  the  difference  of  the  d-c  outputs  is  indicated  on  the  two-winding  zero 
center  meter.  By  the  use  of  the  difference  of  the  sum  and  difference  d-c  outputs  the  indi- 
cation obtained  is  very  closely  proportional  to  Aft  —  Aft  from  0  to  ±180°  and,  as  shown 

above,  is  therefore  also  proportional  to  (  -r-^ IT  I  »  *ke  envelope  delay  distortion  in 

\oto          uco/ 

microseconds.  If  the  area  under  the  plotted  curve  is  integrated  step  by  step  by  means 
of  a  planimeter  or  graphical  methods,  the  phase  vs.  frequency  characteristic  can  also  be 
obtained  which  is  the  form  of  data  required  for  use  in  designing  phase-correcting  networks. 

By  the  introduction  of  a  motor-driven  signal  generator  and  interlocking  arrangements 
at  the  sending  end  and  avc  amplifiers  in  place  of  the  attenuators  and  a  recording  meter 
at  the  receiving  end,  this  circuit  can  be  adapted  to  automatic  recording  of  both  the  attenu- 
ation and  envelope  delay  distortion  vs.  frequency  characteristics. 

After  the  initial  phase  correction  is  made  by  means  of  phase  correctors  in  the  line,  the 
residual  delay  distortion  is  considerably  reduced  and  it  is  necessary  to  increase  the  delay 
sensitivity  of  the  measuring  circuit  to  obtain  data  for  more  accurate  phase  equalization; 
this  is  accomplished  by  insertion  of  harmonic  multipliers  at  A.  Similarly  the  phase  sensi- 
tivity may  also  be  increased  by  increasing  the  frequency  interval,  33.  The  objection  to 
this  is  that  it  does  not  catch  the  narrow  interval  variations. 

16.  NOISE  MEASUREMENTS 

Telephone  circuit  currents  other  than  those  produced  by  acoustic  pressures  on  the 
transmitters  (e.g.,  currents  produced  by  electromagnetic  or  electrostatic  induction  from 
power  circuits  or  from  other  telephone  circuits,  currents  produced  by  thermal  noise, 
vacuum-tube  noise,  defective  components,  etc.)  produce  noise  in  a  telephone  receiver 
connected  to  the  circuit,  these  currents  being  called  noise  currents.  Not  all  frequency 
components  of  noise  have  the  same  interfering  effect  on  a  telephone  conversation,  since 
the  human  ear  and  the  telephone  system  do  not  respond  equally  to  all  frequencies.  '  (See 
also  Coordination  of  Communication  and  Power  Systems,  Section  10.)  A  measurement  of 
the  total  noise  power  on  a  telephone  circuit  would,  therefore,  not  be  a  true  indication  of 
its  interfering  effect. 

For  many  years  noise  currents  were  measured  by  comparing  the  actual  noise  as  heard 
in  a  receiver  to  an  adjustable  standard  noise  produced  by  a  buzzer,  the  ear  weighting  the 
different  frequency  components  of  the  noise.  Difficulty  in  comparing  the  noise  with  the 


CROSS-TALK  MEASUREMENT 


11-37 


standard  when  it  did  not  have  the  same  frequency  components,  and  variations  between 
different  observers  when  measuring  the  same  noise,  resulted  in  the  development  of  meter 
methods,  thejdevice  for  measuring  telephone  circuit  noise  being  known  as  a  circuit  noise 
meter  or  noise-measuring  set  (see  references  13  and  14).  This  consists  of  a  high-gain 
amplifier  a  weighting-network,  a  rectifier,  and  a  d-c  meter  as  shown  in  Fig.  7. 


600o> 


FIG.  7.     Circuit  Noise  Meter 

A  typical  weighting  network  and  amplifier  together  have  the  response  characteristic  as 
shown  in  Fig.  8.  This  curve  is  based  on  a  large  amount  of  experimental  data  and  takes 
into  account  the  typical  receiver-ear  sensitivity  and  terminal  trunk  transmission  character- 
istics. For  lines  carrying  program,  the  weighting  characteristic  would  be  different,  giving 
more  weight  to  the  upper  and  lower  frequencies. 

The  rectifier  circuit  has  been  designed  so  that  the  d-c  output  for  a  steady-state  complex 
input  to  the  rectifier  is  proportional  to  the  square  root  of  the  sum  of  the  squares  of  the 
individual  single-frequency  voltages  in  the  complex  input.    The  output  is  a  function  of 
the  average  power  impressed  on  the  rectifier  and 
not  of  wave  shape. 

A  limited  db  scale  on  the  meter  and  a  gain  con- 
trol calibrated  in  decibels  are  provided  for  indi- 
cating the  noise  level  in  decibels  above  "reference 
noise,"  which  is  the  term  given  to  any  circuit  noise 
which  would  produce  a  meter  reading  of  zero,  the 
same  reading  as  would  be  produced  by  sending 
10~12  watt  of  1000-cycle  power  into  the  circuit 
noise  meter  (600  ohm  input) . 

Reference  noise  is  defined  in  this  manner  so  as 
to  facilitate  calibration  and  measurement.  For  a 
single  type  of  telephone  instrument  and  corre- 
sponding weighting  network  the  definition  is  suffi- 
cient. However,  there  are  several  instruments 
with  different  frequency-response  characteristics, 
and  each  requires  a  separate  weighting  network. 
The  method  of  calibration  makes  these  networks 
all  give  the  same  reading  for  a  1000-cycle  input, 
but  noise  measured  as  equal  with  different  net- 
works may  not  be  equal  in  interfering  effect  when 
heard  with  the  corresponding  instruments.  In 
practice,  it  is  customary  to  express  noise  magni- 
tudes in  dba  (decibels  adjusted)  by  adjusting  the 
reading  in  db  above  reference  noise  so  that  equal  magnitudes  (in  dba)  represent  equal 
interfering  effects  for  different  types  of  instruments.  A  different  adjustment  is  required 
for  each  type  of  telephone  instrument. 

The  meter  and  associated  circuits  have  a  dynamic  characteristic  such  that  the  response 
to  sounds  of  short  duration  approximately  simulates  that  of  the  ear. 


/ 

"""s 

^ 

10 

/ 

^ 

x 

o 

\ 

X 

1->o 

s 

\ 

£ 

v 

S, 

°  30 

\ 

<D  40 

•3 

=§  5Q 

60 

0             1             2             3            4             E 

Frequency-kilocycles  per  sec 

FIG.     8.     Frequency     Characteristic     o 
Typical  Weighting  Network 

17.  CROSS-TALK  MEASUREMENT 

Transmission  between  separate  communication  circuits  is  called  cross-talk*  When  the 
cross-talk  comes  principally  from  one  other  circuit,  it  is  measured  in  the  same  way  as  a 
transmission  loss  by  sending  testing  power  into  the  disturbing  circuit  and  measuring  the 
cross-talk  power  received  in  the  other,  the  ratio  between  the  powers  being  expressed  in 
decibels,  since,  if  a  circuit  is  not  overloaded,  the  ratio  is  substantially  independent  of  the 
actual  power.  This  type  of  measurement  is, called  a  cross-talk  coupling  measurement. 

In  measuring  the  cross-talk  coupling  between  two  coterminous  two-wire  circuits,  the 
generator  which  supplies  the  testing  power  and  the  receiving  device  which  measures  the 
received  power  are  connected  at  the  same  ends  of  the  circuit  as  in  Fig.  9,  the  measurement 
being  called  a  near-end  measurement.  Four-wire  and  carrier  circuits  have  separate 
transmitting  and  receiving  paths.  For  these  types  of  circuits,  it  is,  therefore,  often  desir- 


11-38 


ELECTRICAL  MEASUREMENTS 


able  to  connect  the  disturbing  generator  and  the  receiving  device  at  opposite  ends  of  the 
circuits,  as  shown  in  Fig.  10,  this  type  of  cross-talk  coupling  measurement  being  called  a 
far-end  measurement. 


Disturbing  Circuit 


Disturbed  Circuit 


FIG.  9.     Near-end  Cross-talk  Coupling  Measurement 


Disturbing  Circuit 


Disturbed  Circuit 


FIG.  10.     Far-end  Cross-talk  Coupling  Measurement 

As  the  frequency  of  the  disturbing  power  is  changed,  the  cross-talk  coupling  between 
two  circuits  varies  over  a  wide  range.  Single-frequency  cross-talk  measurements  are 
therefore  of  little  value,  and  generators  of  complex  wave  shape  are  used  to  obtain  results 
approximating  actual  talking  conditions.  Either  warbler  oscillators  or  tube  noise  genera- 
tors whose  energy-frequency  spectra  are  shaped  to  simulate  the  human  voice  are  used  as 
power  sources.  The  warbler  oscillator  contains  a  frequency-changing  device  that  causes 
the  frequency  to  sweep  over  a  wide  range  several  times  per  second.  The  meter  in  the 
receiving  or  measuring  device  averages  the  results  over  the  range. 

Except  when  there  is  trouble,  the  cross-talk  between  any  two  circuits  is  generally  very 
small.  However,  when  there  are  many  circuits  in  a  group,  as  in  telephone  cables,  each 
may  produce  a  small  amount  of  cross-talk  in  any  one  circuit  so  that  the  total  cross-talk 
in  that  circuit  may  be  noticeable.  Coming  from  so  many  sources,  it  is  usually  unintel- 
ligible and  is  commonly  known  as  "babble."  The  amount  of  babble  varies  with  the 
actual  volume  of  speech  on  the  other  circuits,  being  greatest  at  periods  when  the  greatest 
number  of  circuits  are  in  use. 

Since  the  sources  of  babble  are  numerous*  and  since  the  volume  on  the  disturbing  circuits 
cannot  be  controlled,  the  babble  noise  on  a  circuit  must  be  measured  in  some  such 
manner  as  speech  volume,  or  noise,  the  measurement  being  one  of  power,  rather  than  a 
power  ratio  measurement  as  in  cross-talk  coupling. 

Tests  are  made  by  connecting  a  high-impedance  vacuum-tube  measuring  device  across 

an  idle  circuit.    This  receiving  device  is  similar  to  that  used  in  transmission  level  or  volume 

tests  with  the  exception  that  the  amplifier  does  not  have  a  uniform  frequency  response,  the 

Pu!se  characteristic  of  the  amplifier  being  simi- 

generator  lar  to  that  used  in  noise  measurements. 

j  1 -A-  ^  If  observations  with  this  device  for  short 

' — i — '  periods  during  several  successive  days 

show  no  abnormal  conditions,  the  circuit 
is  considered  satisfactory.  A  high-imped- 
ance device  is  employed  so  that  it  will 
not  interfere  with  normal  use  of  the  circuit. 


Tertntnation 

Transmitted 
|  -       pulse 

I     Reference  line 


18.  ECHO  TESTING  OF  LINES 


Reflections 

from 
Irregularities 


This  method  of  test  is  an  adaptation  of 
radar  methods  and  has  been  used  in  two 
principal    applications:    (1)    location    of 
gross  faults  in  cable  and  open-wire  cir- 
cuits, and  (2)  detection  of  irregularities  of 
coaxial  cable  circuits  used  for  television. 
Figure  11  shows  a  circuit  applicable  to  this  test.    The  fundamental  circuit  is  similar  for 
both  applications,  the  differences  being  in  the  shape  and  length  of  the  transmitted  pulses, 
t&e  reputation  rates,  and  the  sweep  speeds  (see  references  12,  15,  and  16). 


Echo  Indicator 


11-40  ELECTRICAL  MEASUREMENTS 

resistance  and  reactance  curves  are  no  longer  smooth  but  change  periodically,  curve  B 
showing  the  effect  of  an  irregularity  on  the  effective  resistance.  The  reactance  changes 
in  a  similar  manner.  This  effect  is  utilized  in  locating  the  irregularity,  as  there  is  a  relation 
between  the  separation  of  peaks  on  the  curve  and  the  distance  to  the  irregularity. 

When  an  alternating  current  strikes  a  circuit  irregularity  such  as  a  sudden  change  in 
impedance,  some  of  the  current  is  reflected  towards  the  sending  end,  the  amount  so  re- 
flected depending  upon  the  size  of  the  irregularity.  Sometimes  this  reflected  current  aids 
the  current  entering  the  line  and  sometimes  it  reduces  it,  depending  upon  the  distance  to 
the  irregularity  and  the  frequency.  When  the  distance  from  the  sending  end  of  the  line 
to  the  irregularity  is  great  a  larger  number  of  wavelengths  is  included  between  the  two 
points  than  when  the  line  is  short;  consequently  a  smaller  change  in  frequency  is  necessary 
to  add  an  extra  wavelength.  Each  wavelength,  half  of  which  adds  to  and  half  of  which 
subtracts  from  the  original  current,  thereby  decreasing  or  increasing,  respectively,  the 
impedance  of  the  circuit,  causes  a  peak  or  hump  in  the  curve. 

As  an  example  of  the  manner  in  which  the  distance  to  an  irregularity  can  be  deter- 
mined (see  Fig.  12),  let  d  be  the  distance  to  an  irregularity.  The  reflected  current  must 
travel  from  the  sending  end  to  the  irregularity  and  back  again  to  the  starting  point,  so 
that  it  really  travels  twice  the  distance  or  2d.  If  the  length  of  one  wave  is  TFi,  the  total 
number  of  wavelengths  in  the  reflected  current  equals  twice  the  distance  divided  by  the 
wavelength.  Let  the  number  of  waves  be  designated  by  N;  then  N  =  2d/Wi  at  some 
particular  frequency  which  can  be  called  fa.  Assume  that  fa  is  the  frequency  corresponding 
to  one  of  the  peaks  on  the  impedance  curve. 

As  brought  out  in  the  foregoing,  when  the  frequency  has  been  increased  so  that  one 
more  wavelength  is  included  in  the  double  length  of  line,  another  peak  will  be  produced 
in  the  impedance  curve.  Let  this  frequency  be  designated  as  /2.  There  are  now  N  -f  1 
wavelengths  at  a  frequency  fa,  or  N  +  1  =  2d/W$.  The  distance  d  to  the  irregularity  has 
not  changed,  but  the  length  of  one  wave  has  changed  to  some  value  W%. 

As  the  wave  travels  along  a  telephone  line  at  practically  the  same  velocity  at  all  fre- 
quencies, the  wavelength  can  be  expressed  in  terms  of  velocity  and  frequency.  If  an  alter- 
nating current  flows  over  a  circuit  at  some  velocity  V  miles  per  second,  the  length  in  miles 
of  any  one  wave,  W,  is  equal  to  the  velocity  divided  by  the  number  of  waves  per  second, 
or  the  frequency.  In  other  words,  W  —  V/f. 

As  shown  above,  the  number  of  waves  in  the  double  path  of  the  reflected  current  is 
equal  to  N  for  frequency  fa  and  N  -f-  1  for  frequency  h-  In  turn  N  —  2d/Wi  and  N  -f  1 
=  2d/W*.  Since  the  wavelength  equals  the  velocity  in  miles  divided  by  the  frequency, 
the  wavelength  for  any  particular  frequency  such  as  /  equals  the  velocity  divided  by 
that  frequency.  Therefore, 


and 

Substituting  these  values  of  W\  and  Ws  in  the  equations 

N  =  — 

and 

2d_ 

respectively.    Then 

2J/F      24A 

/i      ~    F 
and 


The  frequency  fi  represents  one  peak  on  the  curve,  and  /a  represents  the  next  peak  as  the 
frequency  increases.  From  the  curve  can  be  determined  the  number  of  cycles  difference 
in  the  two  frequencies  representing  adjacent  peaks,  this  difference,  of  course,  being  equal 
to  A  —  fa.  Combining  the  two  equations  above  so  that  the  term  /2  —  fa  will  be  present, 
subtract  N  from  N  •}-  I,  Then 


D-C  AND  LOW-FREQUENCY  LINE  TESTING 


11-41 


This  gives 


V  = 


V 

*  -  /i) 


From  this  the  distance  to  any  irregularity  may  be  determined  provided  the  difference 
in  frequency  between  two  adjacent  humps  and  the  velocity  with  which  an  alternating 
current  flows  along  the  line  are  known.  The  above  equation  may  be  written  as 


20*  -  /i) 

Expressed  in  words  this  means  that  the  distance  to  any  irregularity  equals  the  velocity 
with  which  an  alternating  current  flows  along  the  line  in  question  divided  by  twice  the 
difference  in  frequency  between  adjacent  humps. 

The  velocity  of  transmission  for  all  types  of  circuits  is  determined  experimentally  by 
introducing  a  known  irregularity  and  solving  the  equation  for  V.  In  practice  it  is  usual 
to  take  the  difference  in  frequency  between  several  adjacent  peaks,  between  approximately 
700  and  1500  cycles,  and  use  the  average  difference  in  velocity  at  different  frequencies 
which  causes  the  interval  between  peaks  to  change  slightly  with  frequencj-. 

Measurements  are  usually  made  with  the  simple  form  of  impedance  bridge  shown  sche- 
matically in  Fig.  12. 


21.  D-C  AND  LOW-FREQUENCY  LINE  TESTING 

Telegraph,  and  telephone  line  test  boards  are  generally  equipped  with  special  types 
of  voltmeters  and  Wheatstone  bridges  for  making  periodic  tests  of  the  line  wires  and  for 
the  location  of  faults.  These  faults  are  of  three  general  types:  grounds,  crosses,  and 
opens;  and  they  may  have  any  value  of  resistance  from  zero  to  several  megohms.  The 
voltmeter  provides  a  simple  method  of  deter- 
mining the  type  and  magnitude  of  a  fault,  as 
shown  in  Figs.  13,  14,  15,  and  16. 


Voltmeter 


Une  pair 


.100,OOO-ohm  voltmeter 


Open  at 
distant  end 


Strap  at 
distaoi  end 


I?  leakage  is  assumed  uniform 

-^-x  length  jn 


Megohms  per  mile 
FIG.  14. 


Insulation  Test 


FIG.  13.     Continuity  Test 


100,000-ohm  voltmeter 


T 


-Open 


.- 

10V 


raegolmrs 


FIG.  15.     High-resistance  Leak  to  Ground 


When  key  is  closed  the  charging  current  will 
cause  a  momentary  deflection.  The  duration  is 
proportional  to  the  capacitance.  Thus  by  com- 
paring the  deflection  obtained  with  the  defective 
wire  with  that  obtained  with  a  good  conductor 
of  known  length  the  distance  to  the  fault  can  be 
estimated. 

FIG.  16.     Voltmeter  Test  for  Open 


For  the  actual  location  of  faults  the  Wheatstone  bridge  is  used  as  illustrated  in  Figs. 
17,  18,  19,  20,  and  21.    In  these  figures  L  is  the  length  of  the  line  and  d  is  the  distance 


Good 


Line 


Strap 


FIG.    17.     Simple   Bridge  to   Measure 
Loop  Resistance.    At  Balance  RL  —  -R. 


FIG.  18.  Simple  Murray  Loop  Test  for  Grounded 
Conductor.  When  A  and  B  are  adjusted  for 
balance 


11-42 


ELECTRICAL  MEASUREMENTS 


from  the  testing  end  to  the  fault.  It  is  assumed  in  the  illustrations  that  the  good  and  faulty 
conductors  used  in  the  bridge  measurement  have  the  same  resistances  per  unit  length  in 
the  case  of  grounds  and  crosses,  and  the  same  capacitances  per  unit  length  in  the  case  of 
opens. 


Strap 


Note:  The  conductor  unit  resistance  will 
not  usually  be  accurately  known.  For  this 
reason  a  fault  is  ordinarily  located  on  a  per- 
centage basis  by  making  both  simple  loop  and 
Varley  measurements. 

FIG.     19.     Simple    Varley    Loop    Test    for 
Grounded  Conductor 

R 


(K  —  ohms  per  conductor  unit  length.) 


When  R  is  adjusted  to  give  minimum 
response  in  telephone  receiver 


FIG.  20.  Simple  Murray  Loop  Test  for 
Opens.  [Suitable  only  for  short  lines  (1/2 
mi  cable).  For  longer  lines  use  Fig.  21. 


>Fault 


Distributed  capadlajoce  I 

^ 

Ry=  resistance  to  balance 
for  capacitance  of 
length  d  of  faulty 
conductor 

Ny=  resistance  to  balance- 
for  resistance  of 
faulty  conductor 


sre    capacitance 
J_of  good  conductor  of 
~  length  L 


to  balance,  for  capacitance 
of  length  L 

ti0=resistajice  to  balance  for  resistance 
of  'good  conductor 


FIG.  21.    To  Locate  an  Open,  a  Bridge  Reading  Is  Taken  on  the  Open  Conductor  and  Compared  with 
the  Reading  Obtained  on  a  Good  Conductor  of  Known  Length 

If  the  fault  is  a  cross  between  wires,  the  second  crossed  wire  is  substituted  for  the  ground 
connection  to  the  battery  key. 

It  is  important  that  the  location  be  as  accurate  as  possible  to  lessen  the  overall  fault 
clearing  time,  and  to  this  end  variations  of  the  simple  bridge  tests  are  used  to  niinimize 
errors  (see  references  22-29). 

BIBLIOGRAPHY 

1.  Bennett,  W.  R.,  Cross-modulation  Requirements  on  Multichannel  Amplifiers  below  Overload, 

Bell  Sys.  Tech.  J.,  Vol.  19,  587-605  (October  1940). 

2.  Kinder,  J.  P.,  Measurement  of  Modulation  in  Carrier  Amplifiers,  Bell.  Labs.  Rec.,  August  1941. 

3.  Tidd,  Rosen,  and  Wenk,  New  Test  Equipment  and  Testing  Methods  for  Cable  Carrier  Systems, 

Trans.  AJ.E.E.,  Vol.  66  (1947). 

4.  Bode,  H.  W.,  Network  Analysis  and  Feedback  Amplifier  Desiffn,  D,  Van  Nostrand.  New  York. 

5.  Black,  H.  S.,  Stabilized  Feedback  Amplifier,  Bell  Sys.  Tech.  J.,  Vol.  13,  1-18  (January  1934). 

6.  AJsberg  and  Leed,  A  Precise  Direct  Reading  Phase  and  Transmission  Measuring  System  for  Video 

Frequencies,  BeU  Sys.  Tech.  J.,  Vol.  28,  231-238  (April  1949). 

7.  Elliot,  J-  S.,  Precise  Measurement  of  Insertion  Phase  Shift,  Bell  Labs.  Rec.,  Vol.  XVI,  No.  8,  285. 

8.  Mead,  S.  P.,  Phase  Distortion  and  Phase  Distortion  Correction,  Bell  Sys.  Tech.  J.,  Vol.  7,  195-224 

(April  192S). 


OVERALL  A-M  RECEIVES  MEASUREMENTS  11-43 

9.  Lane,  C.    E.,   Phase   Distortion  in  Telephone   Apparatus,  Bdl  Sys.  Tech.  J.,  VoL  9,  493-521 
(July  1930). 

10.  Nyquist,  H.,  and  Brand,  S.,  Measurement  of  Phase  Distortion,  Bell  Sys.  Tech.  J.,  Vol.  9,  522-549 

(July  1930). 

11.  Wentz,  J.  F.,  Measuring  Transmission  Speed  of  the  Coaxial  Cable,  Bell  Labs.  Rec.t  June  1939. 

12.  Strieby,  M.  E.,  and  Weis,  C.  L.,  Television  Transmission,  Proc.  I.R.E.,  Vol.  29,  300-321  (July 

1941) ;  also  p.  381. 

13.  Barstow,  J.  M.,  Blye,  P.  W.,  and  Kent,  H.  32.,  Measurement  of  Telephone  Noise  and  Power  Wave 

Shape,  Elec.  Eng.,  Vol.  54,  1307-1315  (December  1935). 

14.  Castner,  T.  G.,  Dietze,  E.r  Stanton,  G.  T.,  and  Tucker,  R.  S,,  Indicating  Meter  for  Measurement 

and  Analysis  of  Noise,  Trans.  AJ.E.E. ,  September  1931,  pp.  1041-1047. 

15.  Schott,  J.  T.,  The  Lookator,  Bell  Labs.  Rec.,  October  1945,  p.  379. 

16.  Abraham,  Lebert,  Maggio,  and  Schott,  Pulse  Echo  Measurements  on  Telephone  and  Television 

Facilities,  Trans.  A.I.E.E.,  Vol.  66,  pp,  541  to  548. 

17.  Swift,  G.,  Amplifier  Testing  by  Means  of  Square  Waves,  Communications,  February  1939. 

18.  Transient  Response  of  a  Broadcast  System,  General  Radio  Experimenter,  April  1940. 

19.  Network  Testing  with  Square  Waves,  General  Radio  Experimenter,  December  1939. 

20.  Williams,  J.T  Square  Wave  Testing  of  Amplifiers,  Radio  JYetrs,  January  1944,  p.  24. 

21.  Ferris,  L.  P.,  and  McCurdy,  R.  G.,  Telephone  Circuit  Unbalances,  J.  AJ.E.E.,  Vol.  SLIII, 

No.  12  (December  1924). 

22.  Palmer,  W.  T.,  A-c  Method  of  Fault-localization  in  Telephone  Cables,  P.O.E.E.  J.,  Vol.  23,  No.  1 

(April  1930). 

23.  Trans.  AJ.E.E.,  Vol.  43,  423  and  1320. 

24  Northrop,  E.  F.,  Methods  of  Measuring  Electrical  Resistance,  McGraw-Hill,  1912. 

25  Ritter,  E.  S.,  Cable  Testing,  Printed  Paper  104,  Institute  of  P.O.E.E.,  1923. 

26.  Edwards,  P.  G.,  and  Herrington,  H.  W.,  The  Location  of  Opens  in  Toll  Telephone  Cables,  Bell  Sys. 
Tech.  J.,  Vol.  6,  No.  1  (January  1927). 

27  Henneberger,  T.  C.,  and  Edwards,  P.  G.,  Bridge  Methods  of  Locating  Resistance  Faults  in  Cable 

Wires,  Bell  Sys.  Tech.  J.,  Vol.  10,  382-407  (July  1931). 

28  Metson,  G.  H.T  An  Accurate  Method  of  Sub-localizing  Cable  Faults,  P.O.E.E.  J.,  VoL  30,  No.  2, 

99  (1937). 

29  Allan,  J.  M.,  Two  Methods  of  Locating  Cable  Faults,  P.O.E.E.  JM  Vol.  29,  Part  2  (July  1946). 


ROUTINE  MEASUREMENTS  ON  A-M  AND  F-M 
BROADCAST  RECEIVERS 

By  W.  O.  Swinyard 

In  receiver  measurements  it  is  sometimes  convenient  to  use  the  logarithm  of  the  stage 
gain  so  that  gains  can  be  added,  instead  of  multiplied,  to  determine  the  overall  non- 
regenerative  gain,  and  to  express  these  in  terms  of  decibels.  It  is  also  convenient  to  discuss 
overall  sensitivities  in  terms  of  decibels  below  1  volt,  in  which  case  the  microvolt  sensitivity 
is  used  to  determine  a  voltage  ratio.  Though  this  is  strictly  a  misuse  of  the  term,  it  is 
convenient  and  will  lead  to  no  serious  difficulty  as  long  as  the  user  is  aware  of  the  limitations 
of  these  practices. 

,  22.  OVERALL  A-M  RECEIVER  MEASUREMENTS 

REQUIRED  TEST  EQUIPMENT.  Standard-signal  generator,  standard  dummy 
antenna,  standard  test  loop,  output  wattmeter,  audio-frequency  generator,  distortion 
meter  or  wave  analyzer,  and  an  auxiliary  1000-kc  signal  generator. 

STANDARD  TEST  CONDITIONS.  Standard  line  voltage  is  117  volts  for  a-c,  d-c,  and 
a-c/d-c  receivers.  Receivers  designed  for  a-c  and  d-c  operation  are  usually  tested  on 
alternating  current  and  check  measurements  are  made  on  direct  current. 

Measurements  on  automobile  receivers  should  be  made  using  a  battery  which  provides 
6.6  volts  at  the  receiver  battery  terminals. 

The  normal  test  voltage  for  receivers  designed  for  farm  lighting  systems  is  36  volts. 

Battery-operated  receivers  should  be  tested  using  new  batteries  of  the  type  and  voltage 
specified  by  the  receiver  manufacturer. 

The  volume  control  and  the  tone  control  or  controls  should  be  set  to  provide  maximum 
400-cycle  output.  If  a  selectivity  control  is  provided,  it  should  be  set,  for  the  initial  tests, 
to  provide  greatest  selectivity.  The  effect  of  the  tone  and  selectivity  controls  on  the  per- 
formance should  be  determined  by  special  tests. 

RECEIVER  ALIGNMENT  CONDITIONS.  The  overall  sensitivity,  selectivity,  and 
range,coverage  of  the  receiver  are  first  measured  without  disturbing  the  receiver  alignment. 
These  measurements  are  followed  by  the  single-stage  measurements,  after  which  the  re- 
ceiver is  aligned,  in  accordance  with  the  manufacturer's  service  instructions  if  available, 
and  complete  overall  measurements  are  made.  Normally,  no  overall  measurements  are 
made  with  the  receiver  aligned  at  each  test  frequency.  However,  in  certain  cases  such 
measurements  might  be  desirable  since  they  would  show  the  effect  of  circuit  misalignment 


11-44 


ELECTRICAL  MEASUREMENTS 


Standard 
dummy 
antenna 

— 

Receiver 

> 

Standard 
dummy 
load 

1 

y- 

400-cycie 
filter 

t- 

Harmonic 
analyzer 

1 

Wattmeter 

FIG.  1,     Arrangement  of  Apparatus  Used  for  Overall  Measurements 


on  the  overall  sensitivity,  assuming  no  regeneration.  In  all  cases,  however,  the  frequencies 
at  which  the  circuits  are  in.  exact  alignment  should  be  noted. 

Figure  I  shows  the  equipment  as  it  is  set  up  for  overall  measurements. 

RANGE  COVERAGE.  The  maximum  and  minimum  frequencies  to  which  the  receiver 
can  be  tuned  in  each  band  are  recorded  for  the  "as  received"  and  later  for  the  aligned 

C°OVERALL  SENSITIVITY  (SENSITIVITY-TEST  INPUT).  This  test  normally  con- 
sists of  determining  the  sensitivity-test  input  at  three  to  six  points  in  each  wave  band. 

The  output  of  the  stand- 
ard-signal generator  is  fed 
into  the  input  terminals  of 
the  receiver  through  the 
standard  dummy  antenna. 
The  output  meter  is  con- 
nected to  the  secondary  of 
the  output  transformer, 
and  the  load  is  adjusted  to 
the  proper  value.  The  re- 
ceiver is  then  turned  on 
and  the  voltage  of  the 
power  source  set  to  the 
correct  value.  After  it 
has  throughly  warmed  up 
it  is  tuned  to  the  modu- 
lated signal-generator  out- 
put and  the  controls  are  adjusted  to  provide  maximum  400-cycle  output.  The  400-cycle 
filter  is  then  switched  in  to  remove  the  noise,  and  the  sensitivity-test  input  is  determined. 
It  may  be  measured  in  decibels  below  1  volt  or  in  microvolts. 

If  the  receiver  employs  a  selectivity  control  its  effect  on  the  sensitivity  is  determined 
by  additional  measurements  which  are  usually  made  at  600  kc,  1000  kc,  and  1400  kc. 

The  sensitivity  of  battery-operated  receivers  is  measured  in  the  normal^way  and  with 
A  and  B  batteries  whose  terminal  voltage  has  dropped  to  1.1  volts  for  the  A  and  to  60 
per  cent  of  the  nominal  value  for  the  B  battery. 

A  400-cycle  filter  used  with  the  output  wattmeter  usually  will  reduce  the  noise  voltage 
in  the  output  to  a  negligible  level.  However,  there  may  be  cases  where  noisy  or  extremely 
sensitive  receivers  are  being  measured  when  allowance  must  be  made  for  residual  noise. 
EQUIVALENT-NOISE-SIDE-BAND  INPUT.  The  equivalent-noise-side-band  input  is 
taken  equal  to  the  input  of  a  single  side  band  of  400-cycle  modulation  which  will  produce 
an  output  from  the  receiver  equal  to  the  noise  output,  other  conditions  being  the  same. 
The  reason  noise  of  both  side  bands  has  to  be  identified  with  a  single-side-band  component 
of  modulation  is  that  there  is  a  random-phase  relation  between  the  noise  side  bands  as 
distinguished  from  the  specific  phase  relations  existing  between  the  carrier  and  each  pair 
of  side-band  components  of  modulation.  The  equation  for  JEN  SI  is: . 

1?  ' 

•En  —  0.32?$  ~^~t 
•&* 

where  En  is  the  desired  ENSI  in  microvolts,  E8  is  the  carrier  level  in  microvolts,  Enr  is 
the  rms  output  voltage  of  the  noise  alone  with  an  unmodulated  carrier  applied  ?to  the 
antenna,  and  Esf  is  the  output  voltage  due  to  modulating  this  carrier  30  per  cent  at  400 
cycles.  The  coefficient  0.3  is  required  because  the  test  input  is  modulated  30  per  cent,  and 
only  one  side  band  is  considered. 

To  obtain  ENSI,  the  sensitivity  is  first  determined  making  due  allowance  for  noise. 
This  gives  a  convenient  value  for  E8  in  microvolts.  The  corresponding  400-cycle  output 
voltage  gives  the  value  of  Eaf.  The  modulation  is  then  removed,  the  400-cycle  filter 
switched  out,  and  the  rms  voltage  due  to  the  noise  is  noted.  This  gives  the  value  of  En' 
if  a  thermocouple  meter  is  used  or  if  the  proper  correction  is  applied  for  the  particular 
meter  used.  The  corresponding  ENSI  may  then  be  computed  from  the  equation  given 
above. 

The  value  of  E/  and  En'  may  be  observed  at  a  higher  signal  input  level  since  only  their 
ratio  enters.  A  good  practice  is  to  increase  the  carrier  input  at  which  ENSI  measurements 
are  made,  whenever  the  ratio  En'/E8f  exceeds  1 .  This  makes  the  effective  increase  in  noise 
power  due  to  beats  between  noise  side  bands  negligible  relative  to  the  power  due  to  beats 
between  the  noise  side  bands  and  the  carrier. 

It  should  be  noted  that  it  is  possible  to  determine  ENSI  without  using  a  400-cycle 
filter.  Where  this  procedure  is  followed,  it  is  convenient  to  set  the  attenuator  to  the  point 


OVERALL  A-M  RECEIVER  MEASUREMENTS  11-45 

which  makes  the  noise-output  power  equal  to  the  signal-output  power.  ENSI  is  then  the 
signal  input  multiplied  by  the  per  cent  modulation. 

SELECTANCE  RATIOS.  It  is  of  interest  to  know  how  well  the  receiver  discriminates 
against  signals  located  10  kc  away  on  either  side  of  the  desired  channel.  The  regular  overall 
sensitivity  setup  is  used.  The  selectance  ratio  in  decibels  is  the  difference  between  the 
.signal  input  in  decibels  below  1  volt  required  for  normal  test  output  on  the  channel  to 
which  the  receiver  is  tuned  and  that  required  for  normal  test  output  on  the  adjacent 
channel.  The  selectance  ratios  are  usually  measured  at  600  kc,  800  kc,  1000  kc,  and  1400 
kc  in  the  broadcast  band  and  at  corresponding  points  in  the  long-wave  band  if  one.  is  used. 

IMAGE  RATIOS.  The  setup  for  these  measurements  is  the  same  as  that  for  overall 
sensitivity.  After  measuring  the  sensitivity,  the  receiver  tuning  is  left  undisturbed  and 
the  signal  generator  is  set  to  the  image  frequency.  The  least  signal-input  voltage  with  the 
signal  modulated  30  per  cent  at  400  cycles  required  for  normal  test  output  is  the  image 
sensitivity.  The  image  ratio  in  decibels  is  the  difference  between  the  overall  and  image 
sensitivities,  when  both  are  expressed  in  decibels  below  1  volt. 

There  are  other  spurious  responses  such  as  the  half  i-f  image  which  are  due  to  oscillator 
harmonics  beating  against  either  the  fundamental  of  an  interfering  signal  or  harmonics 
of  that  signal  which  may  be  generated  in  the  receiver.  However,  these  are  not  usually 
measured  in  a  routine  receiver  analysis. 

I-F  REJECTION  RATIOS.  The  sensitivity  of  the  receiver  to  the  intermediate  fre- 
quency is  determined  with  the  regular  setup  used  for  overall-sensitivity  measurements 
(including  the  standard  "all-wave"  dummy  antenna).  The  i-f  rejection  ratio  in  decibels 
is  the  difference  between  overall  and  i-f  sensitivities  in  decibels  below  1  volt.  Measure- 
ments are  usually  made  at  three  points  in  each  wave  band.  In  the  broadcast  band  these 
points  are  usually  the  two  lowest  and  the  highest  frequency  test  points. 

OVERALL  SELECTIVITY.  The  overall  selectivity  is  measured  at  the  center  of  the 
band  (1000  kc  in  broadcast  receivers).  The  setup  is  the  same  as  that  used  for  overall- 
sensitivity  measurements,  and  the  measurement  can  conveniently  be  made  after  the 
measurement  of  sensitivity  and  ENSI  at  1000  kc.  The  total  widths  of  the  selectivity 
curve  at  points  6,  20,  40,  60,  and  80  db  down  from  the  resonant  voltage  peak  are  recorded 
in  kilocycles  as  We,  W%»  1^40,  Weo,  and  W&.  The  procedure  is  as  follows:  the  receiver  is 
tuned  to  resonance  at  1000  kc  and  the  attenuator  set  to  the  sensitivity-test  input;  the 
signal  generator  is  then  tuned  off  resonance  and  the  output  voltage  is  increased  6  db;  the 
generator  is  tuned  toward  resonance  until  normal  test  output  is  secured  and  the  frequency 
is  noted;  the  generator  is  then  tuned  through  resonance  to  the  point  on  the  other  side  where 
normal  test  output  is  secured  and  the  frequency  is  again  noted.  The  difference  in  kilo- 
cycles between  these  two  readings  is  W&  For  WM,  W&,  etc.,  the  process  is  repeated.  At 
this  point  it  should  be  pointed  out  that  the  modulation  should  be  removed  while  the 
generator  is  tuned  through  resonance  to  avoid  damage  to  the  output  meter  and  that  care 
should  be  taken  to  avoid  errors  in  the  measurement  due  to  possible  back-lash  in  the  signal 
generator  frequency-dial  mechanism. 

Where  variable  i-f  selectivity  is  employed,  selectivity  measurements  are  usually  made 
at  each  position  of  the  selectivity  control  switch,  or  at  three  positions,  in  the  case  of 
continuously  variable  selectivity,  at  the  extremes  and  middle  settings  of  the  control. 

An  additional  measurement  of  selectivity  is  made  in  the  case  of  battery-operated  re- 
ceivers, namely,  under  the  ''dead  battery"  conditions  previously  described. 

For  a  receiver  employing  a  long-wave  band,  it  is  desirable  to  measure  the  selectivity  at 
the  midpoint  of  this  band  since  the  normal  selectivity  of  low-frequency  preselector  circuits 
may  result  in  appreciable  narrowing  of  the  skirt  selectivity. 

In  the  case  of  a  receiver  incorporating  a  double  avc  system  normal  selectivity  measure- 
ments will  indicate  sharper  selectivity  than  is  actually  provided  by  the  selectivity  of  the 
resonant  circuits.  This  is  true  to  some  extent  in  any  receiver  which  has  less  selectivity  to 
the  avc  detector  than  to  the  signal-frequency  detector.  In  some  cases  it  may  be  advisable 
to  disable  the  avc  before  making  selectivity  measurements. 

WHISTLE  MODULATION.  When  a  signal  having  a  frequency  close  to  twice  the 
intermediate  frequency  is  being  received  it  is  often  accompanied  by  an  audible  whistle  or 
tweet.  This  is  measured  at  several  values  of  signal-input  level  and  compared  with  the 
corresponding  400-cycle  modulated  signal  by  expressing  the  whistle  in  terms  of  the  per 
cent  modulation  at  400  cycles  required  to  give  an  output  voltage  equal  to  that  resulting 
from  the  whistle. 

Measurement  of  the  equivalent  whistle  modulation  is  made  with  the  regular  setup  for 
overall-sensitivity  measurements.  The  signal  generator  with  the  output  set  to  the  appro- 
priate value  and  with  zero  modulation  is  tuned  to  approximately  twice  the  intermediate 
frequency.  The  exact  setting  is  chosen  by  slowly  turning  the  generator  from  about  30  kc 
below  to  30  kc  above  the  frequency  that  is  twice  the  intermediate  frequency  while  the 


11-46  ELECTRICAL  MEASUREMENTS 

receiver  timing  control  is  rocked.  The  generator  is  set  at  the  point  providing  the  greatest 
whistle  output.  The  receiver,  not  the  generator,  is  detuned  to  produce  a  whistle  of  max- 
imum intensity  to  the  ear,  the  volume  control  being  set  so  that  the  output  voltage  at  this 
point  is  well  below  the  overload  level.  The  400-cycle  filter  is  then  switched  in;  the  modula- 
tion is  applied  to  the  signal  generator  and  adjusted  until  the  output  approaches  as  closely 
as  possible  the  output  previously  noted.  The  ratio  of  the  400-cycle  output  voltage  to  the. 
whistle  voltage  multiplied  by  the  per  cent  modulation  gives  the  whistle  modulation  in  per 
cent.  This  procedure  is  followed  for  inputs  of  0,  20,  40,  and  60  db  below  1  volt. 

The  above  procedure  must  be  modified  when  measuring  the  whistle  modulation  at  input 
levels  of  80  db  below  1  volt  and  less,  since  the  noise  tends  to  mask  the  whistle  when  the 
filter  is  not  used.  Therefore,  the  400-cycle  filter  is  switched  in  and  the  whistle^  carefully 
adjusted  to  give  maximum  output.  Then  the  receiver  is  tuned  slightly  to  one  side  of  the 
carrier,  the  modulation  applied,  and  the  whistle  modulation  determined  as  before.  The 
equivalent  whistle  modulation  measured  at  400  cycles  is  usually  substantially  less  than  that 
of  higher  frequency  whistles,  when  appreciable  avc  voltage  is  developed.  However,  these 
low-level  measurements  are  usually  only  important  qualitatively,  and  they  indicate  the 
need  for  work  on  the  receiver  to  remove  their  causes.  In  thoroughly  shielded  receivers  of 
correct  design  the  whistle  is  usually  not  measurable  with  inputs  below  80  db  below  1  volt. 

OUTPUT  AND  AVC  CHARACTERISTICS.  These  measurements  are  made  in  the 
middle  of  the  band  (1000  kc  for  broadcast  receivers)  using  the  setup  and  test  conditions 
for  overall  sensitivity.  The  signal  is  modulated  successively  at  0,  10,  and  30  per  cent  for 
each  value  of  signal  input.  The  signal  input  is  varied  from  120  db  to  0  db  below  1  volt, 
and  the  audio  output  for  each  of  the  three  modulation  percentages  is  plotted  against  the 
corresponding  signal  input.  The  400-cycle  filter  is  switched  in  to  remove  the  noise  from 
that  portion  of  the  30  per  cent  curve  which  is  below  the  overload  level,  and  the  data  are 
recorded  with  and  without  the  filter  for  this  portion  of  the  curve.  The  0  per  cent  output 
curve  indicates  the  noise  output  of  the  receiver  at  full  sensitivity.  It  also  indicates  the 
presence  of  hum  modulation  and  motorboating  at  high  signal-input  levels. 

The  avc  characteristic  is  taken  in  the  same  manner  as  the  30  per  cent  modulation  curve 
except  that  the  volume  is  reduced  sufficiently  to  prevent  overloading  the  output  amplifier. 
Usually  a  reduction  in  the  output  voltage  of  6  db  with  a  0  db  below  1  volt  signal  input  is 
sufficient.  The  400-cyele  filter  should  be  used  where  a  measurable  amount  of  noise  is 
present. 

AVC  FIGURE  OF  MERIT.  The  avc  figure  of  merit  can  be  obtained  from  the  avc  char- 
acteristic. It  is  the  number  of  decibels  decrease  in  signal  input  necessary  to  reduce  by  10 
db  the  output  obtained  at  a  signal  input  level  of  20  db  below  1  volt. 

TWO-SIGNAL  SELECTIVITY  (CROSS-TALK  INTERFERENCE).  For  these  meas- 
urements the  generators  may  be  coupled  to  the  receiver  under  test  in  either  a  series  or 
parallel  arrangement.  The  measurement  is  made  at  1000  kc  for  two  values  of  desired  signal 
input:  46  db  and  0  db  below  1  volt  for  home  receivers  and  46  db  and  14  db  below  1  volt 
for  automobile  receivers.  The  1000-kc  signal  is  tuned  in  with  the  volume  control  set  to 
provide  a  medium-strength  a-f  output  from  a  1  per  cent  modulated  signal.  Modulation 
is  then  removed;  the  two  generators  are  connected  together  with  the  interfering  signal 
modulated  30  per  cent  and  introduced  at  amplitudes  which  result  in  the  output  previously 
noted  for  the  1  per  cent  modulated  desired  signal.  The  interfering  signal  measurements 
are  taken  at  every  channel  on  both  sides  of  1000  kc  up  to  a  100-kc  difference  unless  the 
necessary  signal  input  reaches  the  maximum  output  of  the  generator  before  the  plus  and 
minus  100-kc  points  have  been  reached. 

If  an  auxiliary  1000-kc  generator  is  used,  the  necessary  output  adjustments  can  be  made 
using  the  standard-signal  generator  modulation  for  both  desired  and  interfering  signals. 
When  the  two  are  connected  in  series  for  the  test  the  auxiliary  1000-kc  generator,  having 
no  modulation,  serves  as  the  desired  signal. 

It  should  be  noted  that  the  impedances  of  the  two  standard  dummy  antennas  which  are 
necessary  if  the  signal  generators  are  connected  in  parallel  should  be  double  the  normal 
values.  If  the  two  generators  have  equal  output  impedances  independent  of  the  attenuator 
settings,  the  effective  output  voltage  produced  by  either  one  is  only  half  the  indicated 
output. 

In  making  the  two-signal  selectivity  tests  two  whistles  will  usually  be  encountered  on 
the  low-frequency  side  of  1000  kc.  If  the  intermediate  frequency  is  455  kc,  the  first  occurs 
at  approximately  955  kc  and  is  due  to  the  beat  between  the  fundamental  of  the  receiver 
oscillator,  1455  kc,  and  the  second  harmonic  of  the  input  signal.  The  second  occurs  at 
approximately  910  kc  and  is  the  regular  twice-i-f  whistle  to  which  previous  reference  has 
been  made. 

HARMONIC  DISTORTION.  A  distortion  analysis  is  made  at  1000  kc  using  the  setup 
for  overall  sensitivity  in  conjunction  with  a  distortion  meter  or  a  wave  analyzer.  Three 


OVERALL  A-M  RECEIVER  MEASUREMENTS  11-47 

sets  of  measurements  of  per  cent  harmonics  are  made.  For  the  first  the  signal  input  is 
maintained  constant  at  46  db  below  1  volt,  with  30  per  cent  modulation  at  400  cycles. 
Harmonics  are  measured  at  several  output  levels  up  to  and  including  the  maximum  obtain- 
able. It  is  desirable  to  choose  one  output  level  so  that  10  per  cent  total  distortion  results 
since  this  is  usually  chosen  as  representing  the  maximum  undistorted  output.  For  the 
second  set  of  measurements  the  signal  input  is  maintained  constant  at  46  db  below  1  volt 
and  the  receiver  volume  control  is  adjusted  for  normal  test  output.  The  modulation  is 
then  set  to  10  per  cent,  50  per  cent,  and  SO  or  100  per  cent,  and  the  harmonics  are  measured 
for  each  value  of  modulation  percentage.  The  final  measurements  are  made  keeping  the 
modulation  at  30  per  cent,  the  output  at  normal  test  output,  setting  the  signal-input  level 
to  60  db,  40  db,  20  db,  and  0  db  below  1  volt  and  measuring  the  harmonics  at  each  value 
of  signal  input.  These  values  of  signal  input  may  have  to  be  modified  for  receivers  of 
special  types.  If  a  wave  analyzer  is  used  the  total  per  cent  harmonic  distortion  is  calculated 
as  the  square  root  of  the  sum  of  the  squares  of  the  individual  per  cent  harmonics. 

OVERALL  ELECTRICAL  FIDELITY,  The  regular  avera]l-sensitivity-measurement 
setup  is  used  for  this  measurement  except  that  the  signal  generator  is  modulated  30  per 
cent  by  an  a-f  signal  generator  of  variable  frequency  and  the  400-eyele  filter  is  switched 
out.  The  receiver  is  first  tuned  to  1000  kc  using  a  weak  signal.  The  signal  input  is  then 
set  to  46  db  belo\v  1  volt,  and  the  receiver  volume  control  is  set  to  provide  an  output  well 
below  overload.  The  modulation  frequency  is  then  increased  until  the  output  is  reduced 
by  about  14  db,  and  the  tuning  control  is  finally  adjusted  for  minimum  output  at  this 
frequency.  The  modulation  is  then  set  back  to  400  cycles  and  a  final  adjustment  of  the 
output  voltage  is  made.  The  output  is  then  measured  at  a  sufficient  number  of  modulation 
frequencies  to  permit  plotting  a  curve  showing  the  relative  response,  in  decibels,  versus 
modulation  frequency  using  the  400-cycle  output  voltage  as  a  reference.  Curves  are 
made  showing  the  maximum  overall  fidelity  and  the  effect  of  the  tone  control  on  the 
fidelity.  Usually  two  curves  are  recorded:  (1)  with  the  tone  control  set  ics  maximum  highs, 
and  (2)  with  the  tone  control  set  for  minimum  highs,  In  the  case  of  1.4r-volt  battery- 
operated  receivers  the  fidelity  is  also  measured  using  "dead  batteries."  If  a  bass-com- 
pensated volume  control  is  used,  measurements  should  be  made  with  the  arm  set  both 
above  and  below  the  tap  to  show  the  effect  of  the  bass  compensation.  It  may  be  better 
to  disconnect  the  network  from  the  tap  in  order  to  avoid  overload. 

TESTS  ON  PUSH-BUTTON  TUNERS.  Receivers  provided  with  a  mechanical  push- 
button tuning  mechanism  are  subjected  to  tests  devised  to  show  how  accurately  the  push 
buttons  can  be  set  and  how  well  they  return  to  the  frequency  to  which  they  are  set.  For 
these  tests  the  buttons  are  all  set  up  to  tune  a  signal  near  the  high-frequency  end  of  the 
band  to  zero  beat  with  a  signal  from  another  generator  which  is  set  to  the  intermediate 
frequency.  For  each  push  button  the  gang  is  opened,  the  button  actuated,  and  the  re- 
sultant beat  recorded  as  a  frequency  error.  This  is  repeated  several  times,  usually  eight, 
and  similar  tests  are  made  from  the  closed  position  of  the  gang.  Finally  tests  are  made 
with  the  gang  alternately  opened  and  closed  before  the  button  is  pushed.  All  measure- 
ments for  each  button  (24)  are  then  averaged  taking  account  of  the  sign,  and  the  result  is 
recorded  as  a  setting  error.  The  setting  errors  for  each  button  are  averaged  without 
regard  to  sign  to  secure  the  average  setting  error.  The  deviations  of  all  the  errors  for  each 
button  from  the  figure  represent  ing  the  setting  error  for  that  button  averaged  without 
regard  to  sign  give  the  mean  tuning  deviation.  The  average  mean  tuning  deviation  is 
the  average  of  the  mean  tuning  deviations. 

MEASUREMENTS  USING  STANDARD  TEST  LOOP.  The  following  measurements 
which  employ  the  standard  test  loop  (Fig.  2)  are  made  on  loop  receivers:  sensitivity, 

Tubular  shield 
10'7in  diameter 


_£03-ohm  resistor 
hoosktg 


FIG.  2.    Test  Loop 


11-48  ELECTRICAL  MEASUREMENTS 

JSNST  image,  i-f  and  selectance  ratios,  and  loop  figure  of  merit.  The  test  loop  is  connected 
to  the  output  of  the  signal  generator  and  set  up  24  in.  away  from  the  receiver  loop  with 
the  loops  arranged  coaxially.  The  overall  sensitivity  in  decibels  below  1  volt  per  meter  is 
20  db  below  the  signal  generator  attenuator  reading  when  the  input  is  adjusted  for  normal 
test  output.  The  loop  figure  of  merit  is  the  difference  between  the  sensitivity  in  decibels 
below  1  volt  per  meter  and  the  sensitivity  at  the  first  grid  measured  in  decibels.  It  is 
recorded  as  minus  when  the  absolute  magnitude  of  the  overall  sensitivity  in  decibels  below 
1  volt  per  meter  is  less  than  the  first  grid  sensitivity  in  decibels.  Where  no  external 
antenna  connection  is  provided  other  measurements  than  these  must  of  necessity  be  made 
using  the  loop  input.  In  all  cases  the  same  procedure  is  followed  as  has  been  previously 
outlined  except  that  the  input  is  specified  in  decibels  below  1  volt  per  meter.  Many  signal 
generators  operated  with  the  standard  test  loop  will  not  provide  an  input  to  the  receiver 
loop  of  more  than  20  db  below  1  volt  per  meter  when  the  test  loop  is  operated  under  normal 
conditions. 

AUDIO  FEEDBACK  FACTOR.  If  the  audio  amplifier  of  the  receiver  employs  negative 
feedback  a  measurement  of  overall  a-f  gain  is  made  with  the  feedback  removed,  and  the 
increase  in  gain  measured  in  decibels  at  400  cycles  is  expressed  as  the  audio  feedback  factor. 

OSCILLATOR  DRIFT.  This  test  may  be  carried  out  as  follows:  The  receiver  is  tuned 
to  a  secondary  frequency  standard  or  to  a  signal  generator  of  known  stability  character- 
istics. The  resultant  beat  produced  as  the  oscillator  drifts  is  measured  by  zero  beating  it 
against  an  audio  signal  from  an  amplifier  fed  by  an  a-f  generator.  Oscillator  drift  measure- 
ments are  made  at  some  point  near  the  high-frequency  end  of  each  band.  The  receiver 
is  tuned  to  about  the  right  point  and  then  turned  on  and  as  quickly  as  possible  tuned  to 
zero  beat  against  the  frequency  standard  or  signal  generator.  Measurements  are  made 
every  5  minutes  for  a  half  hour  and  then  every  15  minutes  for  2  hours  or  until  the  oscillator 
frequency  has  become  stabilized.  If  the  oscillator  drift  is  such  that  the  beat  note  goes  above 
the  range  of  audibility  the  drift  can  be  determined  directly  by  zero  beating  with  the 
signal  generator  for  each  measurement.  The  direction  of  drift  can  be  determined  by  noting 
the  effect  on  the  beat  note  of  a  small  change  in  the  capacitance  in  the  oscillator  tank 
circuit — a  change  such  as  is  produced  by  bringing  a  finger  up  close  to  the  tuning  condenser 
stator. 

23.  SINGLE-STAGE  MEASUREMENTS 

REQUIRED  TEST  EQUIPMENT.  Standard-signal  generator,  standard  dummy 
antenna,  output  wattmeter,  vacuum-tube  voltmeter,  tuning  wand,  reactance  meter  or  Q 
meter,  high-resistance  voltmeter,  and  a  wattmeter. 

HIGH-FREQUENCY  MEASUREMENT  PRECAUTIONS.  In  many  cases  measure- 
ments cannot  be  made  properly  by  substituting  a  vacuum-tube  voltmeter  for  a  tube  or 
by  placing  it  in  parallel  with  a  tube.  The  inherent  capacitance  of  the  stage  may  be  so  low 
that  a  change  in  its  value  may  decidedly  influence  the  shape  of  the  selectivity  curve.  In 
such  cases  it  is  suggested  that  the  procedure  to  be  described  later  for  use  in  making  single- 
stage  measurements  on  f-m  receivers  be  followed. 

ANTENNA  GAIN  AND  BAND  WIDTH.  The  voltage  gain  from  the  antenna  to  the 
first  grid  is  measured  at  all  the  test  frequencies  used  for  sensitivity  measurements.  The 
width  of  the  resonance  curve  6  db  down  is  measured  at  three  points  in  each  wave  band. 
For  these  measurements  the  signal  generator  output  voltage  unmodulated  is  fed  into  the 
receiver  antenna  circuit  through  the  standard  dummy  antenna.  The  voltage  developed 
across  the  output  of  the  antenna  circuit  is  measured  by  means  of  a  vacuum-tube  voltmeter 
to  which  the  lead  normally  connected  to  the  grid  of  the  first  tube  is  connected.  The  signal 
generator  output  voltage  is  increased  by  6  db  and  the  generator  is  then  detuned  on  each 
side  of  resonance  until  the  output  drops  to  the  reference  value.  If  there  are  two  tunable 
circuits  ahead  of  the  first  tube  the  gain  and  band  widths  are  also  measured  on  the  first 
circuit  alone.  Where  measurements  are  made  at  high  frequencies,  the  precautions  previ- 
ously noted  must  be  observed. 

COIL  MISALIGNMENT.  When  there  are  two  tunable  circuits  ahead  of  the  first  tube 
or  an  antenna  and  an  r-f  stage,  the  misalignment  of  these  circuits  can  be  measured  by 
means  of  a  reactance  meter  or  a  Q  meter.  For  a  receiver  with  a  tuned  r-f  stage,  the  re- 
ceiver tuning  condenser  is  set  at  a  minimum  capacitance  and  the  standard  dummy  antenna 
is  connected  across  the  antenna-ground  terminals  of  the  receiver.  The  r-f  circuit  is  used 
as  a  standard  and  connected  across  the  condenser  terminals  of  the  Q  meter,  with  a  standard 
coil  <5overing  the  desired  range  plugged  into  the  coil  terminals.  The  frequency  of  the  Q 
meter  is  set  to  the  high-frequency  alignment  point  and  the  Q  meter  condenser  adjusted 
for  maximum  Q  indication.  The  antenna  circuit  is  then  substituted  for  the  r-f  circuit  and 
ibe  antenna  trimmer  adjusted  to  bring  the  antenna  circuit  into  exact  alignment  with  the 
r-f  circuit  as  indicated  by  a  maximum  reading  on  the  Q  indicator. 


SINGLE-STAGE  MEASUREMENTS  11-49 

After  the  antenna  and  r-f  circuits  have  been  aligned,  the  Q  meter  is  set  to  each  of  the 
test  frequencies  and  at  each  point  the  r-f  circuit  is  connected  to  the  condenser  terminals 
and  the  Q  meter  tuning  condenser  is  adjusted  for  maximum  Q  indication.  The  antenna 
circuit  is  then  substituted  for  the  r-f  circuit  and  the  Q  meter  condenser  is  readjusted  for 
maximum  Q  indication.  The  difference  between  the  two  condenser  settings  is  the  misalign- 
ment correction  in  micromicrofarads  between  the  two  circuits. 

Another  measurement  is  made  using  the  Q  meter  to  determine  the  effect  changing  the 
antenna  capacitance  has  on  the  broadcast  or  long-wave  band  tuned  antenna  circuit.  For 
these  measurements  the  receiver  tuning  condenser  is  set  to  minimum  capacitance  and  the 
condenser  terminals  of  the  Q  meter  are  shunted  across  the  tuned  antenna  circuit.  A 
dummy  antenna,  identical  to  the  standard  dummy  antenna  except  that  the  series  capaci- 
tance element  is  adjustable,  is  set  to  the  standard  value  of  200  pid  and  connected  across 
the  antenna  input  circuit.  The  Q  meter  is  set  to  500  ke,  and  the  Q  meter  condenser  is 
adjusted  for  maximum  Q  indication.  The  antenna  capacitance  is  then  set  to  60  /*/if, 
100  /A/if,  300  /A/if,  500  wf,  and  short  circuit  (infinite  capacitance),  and  the  alignment 
corrections  are  noted  for  each  value  of  antenna  capacitance.  These  measurements  are 
usually  made  at  500  kc,  600  kc,  700  kc,  1000  kc,  and  1400  kc  for  the  broadcast  band  in 
cases  where  a  high-inductance  antenna  primary  is  used.  The  700-kc  and  1000-kc  points 
may  be  omitted  if  the  alignment  is  good.  If  the  antenna  circuit  employs  a  low-inductance 
primary,  resonant  above  1500  kc,  the  test  points  are  usually  600  kc,  1000  kc,  1200  kc, 
1400  kc,  and  1500  kc.  In  this  case  the  1000-ke  and  1200-kc  points  may  be  omitted  if  the 
alignment  is  good  at  1400  kc.  Corresponding  values  are  used  in  the  long-wave  band  if 
one  is  provided.  In  the  case  of  receivers  intended  to  operate  with  antennas  having  char- 
acteristics substantially  different  from  normal,  values  of  antenna  capacitance  appropriate 
to  the  case  are  chosen  instead  of  the  above-mentioned  values. 

ALTERNATIVE  METHOD  OF  MEASURING  ALIGNMENT  OF  PRESELECTOR 
CIRCUITS.  To  remove  the  possibility  of  error  due  to  regeneration  the  avc  line  is  biased 
by  means  of  a  battery,  4  1/2-9  volts  will  usually  be  sufficient.  AH  circuits  are  then  aligned 
in  accordance  with  manufacturer's  service  instructions.  At  each  test  frequency  the  mis- 
alignment of  the  preselector  circuits  can  be  determined  by  means  of  a  "tuning  wand," 
which  consists  of  a  length  of  insulating  rod,  such  as  hard  rubber  or  Bakelite,  to  which  an 
iron-dust  slug  and  a  brass  slug  are  attached,  one  at  each  end.  This  is  held  inside  or  near 
each  preselector  coil.  If  the  output  of  the  receiver  decreases  when  either  the  brass  or  the 
iron  is  brought  near  the  coil,  no  misalignment  exists.  However,  if  either  end  causes  an 
increase  in  output  the  misalignment  is  equal  to  the  decibel  increase  in  sensitivity.  If  the 
increase  is  caused  by  the  brass,  the  coil  inductance  is  too  high;  if  it  is  caused  by  the  iron, 
the  inductance  is  too  low. 

R-F  STAGE  MEASUREMENTS.  The  voltage  gain  from  the  grid  of  each  r-f  amplifier 
stage  to  the  grid  of  the  following  stage  is  measured  at  all  the  regular  test  frequencies,  and 
the  6-db  band  width  is  measured  at  three  points  in  each  band  as  discussed  in  the  section 
covering  antenna  circuits.  The  output  of  the  signal  generator  is  fed  into  the  grid  of  the 
tube  incorporated  in  the  stage  being  measured,  and  a  suitable  vacuum-tube  voltmeter  is 
connected  to  the  lead  which  normally  is  connected  to  the  following  grid.  When  the  signal 
generator  is  connected  so  as  to  replace  a  grid  circuit  which  normally  returns  to  the  avc 
string  it  should  be  connected  through  a  condenser,  with  a  grid  leak  returning  to  the  avc 
circuit  so  as  to  maintain  normal  bias  on  the  tube.  The  grid  leak  is  returned  to  the  avc 
circuit  even  though  the  avc  may  eventually  return  to  ground,  since  otherwise  the  measure- 
ments may  be  in  error  owing  to  the  initial  avc  voltage  caused  by  space  current  due  to  the 
emission  velocity  of  electrons  from  the  diode  cathode. 

I-F  MEASUREMENTS.  Diode  Stage.  The  diode  stage  gain  is  measured  by  connect- 
ing the  vacuum-tube  voltmeter  from  the  diode  plate  to  ground,  thus  not  appreciably 
disturbing  the  loading  on  the  diode  transformer.  The  signal  generator  output  is  connected 
through  the  previously  mentioned  condenser-grid  leak  combination,  to  the  last  i-f  grid, 
and  the  stage  is  aligned  for  maximum  output.  A  measurement  of  the  voltage  gain  of  the 
stage  is  then  made,  and  the  bandwidths  6  db  and  20  db  down  are  determined. 

I-F  AMPLIFIER  STAGE  GAIN  AND  BANDWIDTHS.  Measurements  of  gain  and 
bandwidths  of  i-f  stages  are  made  as  described  above  except  that  the  vacuum-tube  volt- 
meter is  not  shunted  across  the  grid  circuit  of  the  following  tube;  it  replaces  it.  It  often 
is  advisable  to  isolate  the  vacuum-tube  voltmeter  by  means  of  a  high-resistance  grid  leak 
and  condenser  if  these  elements  are  not  incorporated  in  the  instrument. 

MODULATOR  STAGE  GAIN  AND  BANDWIDTHS.  This  measurement  is  made  in 
the  manner  described  in  the  above  paragraph.  The  signal-frequency  circuits  should  be 
set  to  the  middle  of  the  band  and  the  oscillator  should  be  operating  for  this  measurement. 

CONVERSION  GAIN.  The  connections  for  the  signal  generator  and  vacuum-tube 
voltmeter  used  in  measuring  the  gain  of  the  modulator  stage  are  employed  to  measure 
conversion  gain,  the  only  difference  being  that  the  signal  generator  is  set  to  the  standard 


11-50  ELECTKICAJO  MEASUREMENTS 

test  frequencies  in  each  band,  conversion  gain  being  denned  as  the  ratio  of  rms  voltage 
of  intermediate  frequency  at  the  grid  circuit  terminals  of  the  first  i-f  transformer  to  the 
rms  voltage  of  signal  frequency  applied  at  the  grid  of  the  modulator  tube.  The  avc  system 
should  be  disabled  for  these  measurements. 

SENSITIVITY  ON  MODULATOR  AND  ON  I-F  GRIDS.  The  least  signal  input  to  the 
last  i-f  grid  required  to  produce  normal  test  output  with  the  signal  modulated  30  per  cent 
at  400  cycles  is  called  the  i-f  sensitivity.  A  corresponding  measurement  made  from  the 
modulator  grid  is  called  the  modulator-grid  sensitivity.  The  i-f  circuits  should  be  aligned 
in  both  cases. 

DETECTOR  SENSITIVITY.  The  detector  sensitivity  is  denned  as  the  amplitude  of 
the  30  per  cent  modulated  rms  voltage  which  must  be  applied  to  the  detector  tube  to 
produce  normal  test  output.  It  is  of  course  an  indication  of  the  total  a-f  gain.  It  is  usually 
determined  indirectly  from  the  measured  diode  stage  gain  and  the  sensitivity  on  the  last 
i-f  grid,  the  difference  between  these  two  quantities  being  the  detector  sensitivity  in  decibels 
below  1  volt. 

OSCILLATOR  VOLTAGE.  The  oscillator  voltage  usually  measured  is  the  d-c  voltage 
across  the  oscillator  grid  leak,  as  calculated  from  the  measured  resistance  value  of  the  grid 
leak  and  the  measured  current  flowing  through  it  when  the  oscillator  is  working.  The 
voltage  is  measured  at  each  test  frequency  throughout  the  range  of  the  receiver.  If  the 
oscillator  circuit  necessitates  some  other  method  of  measuring  oscillator  voltage,  the  pro- 
cedure followed  should  be  noted. 

24.  MISCELLANEOUS  MEASUREMENTS  ON  A-M  RECEIVERS 

HUM  VOLTAGE.  The  rms  value  of  the  hum  in  the  output  of  the  receiver  can  be 
determined  by  means  of  the  output  meter,  provided  sufficient  sensitivity  is  available,  or 
by  connecting  the  primary  of  a  voltage  step-up  transformer  across  the  primary  of  the 
output  transformer.  The  voltage  measured  across  the  secondary  of  this  transformer  by 
means  of  a  vacuum-tube  voltmeter  is  divided  by  the  transformer  voltage  ratio  to  get  the 
required  value  of  hum  in  the  output  circuit.  The  voice  coil  should  be  connected  for  this 
measurement.  The  hum  should  be  measured  for  both  the  maximum  and  the  minimum 
settings  of  the  volume  control.  To  prevent  receiver  noise  from  interfering  with  the 
measurement  a  large  condenser  should  be  connected  from  the  plate  of  the  last  i-f  tube  to 
the  B+  line. 

MINIMUM  OUTPUT  SIGNAL.  This  measurement  is  made  to  determine  how  close 
to  zero  output  the  volume  control  can  reduce  the  signal.  The  receiver  is  tuned  to  a  1000-kc 
46-db  below  1  volt  signal  30  per  cent  modulated,  and  the  power  across  the  standard  dummy 
load  is  measured.  The  400-cycIe  filter  should  be  used  to  remove  hum  and  noise. 

CONDENSER  GANG  ALIGNMENT.  In  the  case  of  gangs  having  uniform  sections 
the  corrections  in  micromicrofarads  required  to  bring  the  antenna  and/ or  r-f  sections  into 
alignment  with  the  oscillator  section  are  measured  for  several  settings  of  the  condenser 
gang.  If  these  measurements  exceed  normal  tolerances  the  gang  can  be  knifed  to  make 
the  necessary  changes. 

D-C  VOLTAGES.  The  principal  d-c  potentials  existing  at  the  tube  elements  should  be 
measured  using  a  high-impedance  voltmeter,  preferably  an  electronic  voltmeter,  and  tabu- 
lated. These  measurements  are  made  directly  after  the  "as  received"  measurements  and 
at  standard  line  or  test  voltage. 

POWER  CONSUMPTION.  The  power  consumed  by  the  receiver  at  standard  line  or 
test  voltage  is  measured  by  means  of  a  suitable  wattmeter.  In  the  case  of  battery-operated 
receivers  the  A  and  B  drains  should  be  measured. 

25.  F-M  RECEIVER  MEASUREMENTS 

Measurement  of  such  characteristics  as  antenna  and  r-f  gain  and  selectivity,  i-f  gain 
and  selectivity,  etc.,  is  independent  of  the  type  of  modulation  which  the  receiver  is  designed 
to  receive,  hence  the  methods  described  above  are  applicable.  In  the  case  of  f-m  receivers, 
higher  frequencies  are  involved;  hence,  the  high-frequency  measurement  precautions  to 
be  described  later  must  be  observed. 

EQUIPMENT.  The  equipment  required  for  routine  measurements  on  a-m  receivers 
is  usually  required  for  f-m  receivers,  since  relatively  few  are  designed  solely  for  frequency 
modulation.  Generally,  frequency  modulation  is  one  of  several  frequency  bands  incor- 
porated in  a  receiver.  Only  measurements  such  as  quieting  and  deviation  sensitivity,  etc., 
where  f-m  modulation  of  the  signal  generator  is  required,  necessitate  the  use  of  an  f-m 
signal  generator. 


OVERALL  PERFORMANCE  TESTS  11-51 

DEFINITIONS  OF  TERMS.  See  also  Standards  on  Radio  Receivers,  1947,  I.R.E., 
Methods  of  Testing  Frequency  Modulation  Broadcast  Receivers. 

Standard  Very-high-frequency  Dummy  Antenna.  The  very-high-frequency  dummy 
antenna  is  a  pair  of  resistors,  one  connected  in  series  with  each  terminal  of  the  signal 
generator,  of  such  value  that  the  total  impedance  between  terminals,  including  the  signal 
generator,  is  300  ohms  "balanced  to  ground. 

Standard  De-emphasis  Characteristic.  The  standard  de-emphasis  characteristic  has  a 
falling  response  with  modulation  frequency,  the  inverse  of  the  standard  pre-emphasis 
characteristic,  equivalent  to  that  provided  by  a  simple  circuit  having  a  time  constant  of 
75  microseconds.  The  characteristic  may  be  obtained  by  taking  the  voltage  across  a 
condenser  and  resistor  connected  in  parallel  and  fed  with  constant  current.  The  resistance 
in  ohms  is  equal  to  0.000075  divided  by  the  capacitance  in  farads.  The  standard  de- 
emphasis  characteristic  is  incorporated  in  the  detector  and/or  audio  circuits  of  the  receiver. 

Standard  Test  Frequencies.  The  standard  test  frequencies  used  in  testing  f-m  receivers 
are  88,  98,  and  108  megacycles.  If  more  than  three  frequencies  are  required,  it  is  suggested 
that  93  and  103  megacycles  be  included.  If  only  one  test  frequency  is  needed,  98  mega- 
cycles should  be  used. 

Standard  Test  Modulation,  This  term  refers  to  a  signal  that  is  frequency  modulated  at 
400  cycles  with  a  deviation  of  30  per  cent  of  the  maximum  system  deviation  of  75  kc.  The 
deviation  due  to  standard  test  modulation  is  therefore  22  1/2  kc. 

Maximum-sensitivity  Test  Input.  The  maximum-sensitivity  test  input  is  the  least 
input  signal  of  a  specified  carrier  frequency  having  standard  test  modulation  which,  when 
applied  through  the  verv,-high-frequency  dummy  antenna,  results  in  standard  test  output 
when  all  controls  are  adjusted  for  greatest  sensitivity.  It  may  be  expressed  in  terms  of 
power  in  decibels  below  1  watt,  in  decibels  below  1  volt,  or  in  microvolts.  Generally  it 
is  advisable  to  use  a  400-cycle  filter  to  remove  the  noise  from  the  output. 

Maximum-d  elation-sensitivity  Test  Input,  The  ma^imiirn-deviation-sensitivity  test 
input  is  the  least  input  signal  of  a  specified  carrier  frequency  having  full  rated  system 
deviation  which,  when  applied  to  the  receiver  through  the  very-high-frequency  dummy 
antenna,  results  in  10  per  cent  distortion  in  the  output  when  the  volume  control  is  adjusted 
to  standard  output.  It  is  expressed  in  decibels  below  1  watt,  in  decibels  below  1  volt,  or 
in  microvolts. 

Deviation-sensitivity  Test  Input.  The  deviation  sensitivity  test  input  is  the  minimum 
deviation  at  400  cycles  of  a  carrier  of  60  db  below  1  volt  required  to  give  maximum  undis- 
torted  output  when  all  controls  are  adjusted  for  greatest  sensitivity.  The  deviation  sensi- 
tivity is  expressed  in  kilocycles  or  as  a  percentage  of  maximum  full  system  deviation. 

Quieting-signal-sensitivity  Test  Input.  The  quietrng-signal-serisitivity  test  input  is 
the  least  unmodulated  signal  input  which,  when  applied  to  the  receiver  through  the  very- 
high-frequency  dummy  antenna,  reduces  the  internal  receiver  noise  to  the  point  where  the 
test  output  rises  30  db  when  standard  test  modulation  is  applied  to  the  input  signal,  the 
volume  control  being  adjusted  to  avoid  audio  overload.  It  is  expressed  in  decibels  below 
1  watt,  in  decibels  below  1  volt,  or  in  microvolts. 

26.  OVERALL  PERFORMANCE  TESTS 

The  setup  for  overall  performance  tests  is  the  same  for  f-m  receivers  as  for  a-m  receivers, 
assuming  that  in  each  case  an  appropriate  signal  generator  and  standard  dummy  antenna 
provide  the  test  signal.  For  measurements  on  f-m  receivers  a  resistor  having  a  value  equal 
to  the  input  impedance  for  which  the  receiver  was  designed  may  be  used  as  a  dummy 
antenna.  If  the  receiver  employs  a  balanced  input,  half  the  resistance  is  used  in  each  side. 
The  conditions  set  down  on  p.  11-43  regarding  receiver  alignment  apply  here  also. 

OVERALL  SENSITIVITY.  The  sensitivity-test  input  for  the  maximum  sensitivity, 
the  maximum  deviation  sensitivity,  the  quieting-signal  sensitivity,  and  the  deviation 
sensitivity  should  be  measured  at  3  to  6  frequencies. 

IMAGE  AND  I-F  RATIO S-  The  procedure  for  determining  these  characteristics  is  the 
same  as  for  a-m  receivers.  The  maximum-sensitivity  test  input  should  be  used  in  deter- 
mining the  image  and  i-f  sensitivity. 

OUTPUT  AND  AVC  CHARACTERISTICS  (LIMITER  CHARACTERISTICS).  This 
measurement  corresponds  to  the  output  and  avc  characteristics  of  a-m  receivers,  and  the 
same  procedure  is  followed  in  obtaining  the  data.  In  the  case  of  f-m  receivers,  the  modula- 
tion percentages  used  are  30  per  cent  (22  1/2  kc)  and  10  per  cent  (7.5  kc).  The  output 
power  versus  signal-input  voltage  is  determined  at  98  megacycles  for  maximum  setting 
Of  the  volume  control.  The  limiter  or  avc  characteristic  is  measured  with  &0  per  cent 
modulation  at  a  volume  control  setting  which  gives  6  db  less  than  the  mnTimnm  output. 
The  output  versus  input  for  zero  modulation  is  also  determined. 


11-52 


ELECTKICAL  MEASUREMENTS 


HARMONIC  DISTORTION.  The  measurements  of  harmonic  distortion  required  for 
an  f-m  receiver  are  as  follows:  per  cent  harmonics  versus  per  cent  modulation  and  per 
cent  harmonics  versus  signal-input  level  and  per  cent  harmonics  versus  output. 

OVERALL  ELECTRICAL  FIDELITY.  The  overall  electrical  fidelity  is  measured  at 
98  megacycles  using  a  signal  input  60  db  below  1  volt,  30  per  cent  modulated.  The  curve 
thus  obtained  should  normally  be  a  close  approximation  to  the  standard  de-emphasis  curve 
for  f-m  receivers  assuming  no  pre-emphasis  in  the  signal  generator.  If  the  receiver  is  multi- 
band,  the  effects  of  the  tone  controls  will  be  shown  by  the  fidelity  measurements  on  the  a-m 
bands;  otherwise,  additional  data  should  be  taken  to  show  these  effects  on  the  fidelity. 

I-F  AMPLIFIER  CHARACTERISTICS.  The  overall  selectivity  of  the  i-f  amplifier 
from  the  modulator  grid  to  the  first  limiter  grid  or,  if  limiters  are  not  employed,  to  the 
plate  of  the  detector  driver  tube,  is  measured  after  the  individual  stage  gain  and  selectivity 
have  been  measured.  The  level  at  the  limiter  grid  should  be  that  required  for  quieting 
the  receiver.  Where  avc  voltage  is  applied  to  i-f  amplifier  stages,  the  overall  i-f  amplifier 
selectivity  characteristics  should  be  measured  for  different  levels  of  signal  input  to  the 
modulator  grid  in  order  to  show  the  detuning  effect  due  to  the  change  in  tube  input  capaci- 
tance with  a  change  in  grid  bias.  Typical  input  levels  are  80  db,  60  db,  and  40  db  below  1 
volt.  The  avc  voltage  in  the  circuit  should  be  held  at  the  center  frequency  level  by  means 
of  a  battery  supply. 

27.  SINGLE-STAGE  MEASUREMENTS 

REQUIRED  TEST  EQUIPMENT.  Standard-signal  generator,  standard  dummy 
antenna,  output  wattmeter,  vacuum-tube  voltmeter,  tuning  wand,  high-resistance  volt- 
meter, and  wattmeter. 

HIGH-FREQUENCY  MEASUREMENT  PRECAUTIONS.  It  will  usuaUy  be  found 
impracticable  in  making  stage  gain  measurements  on  f-m  receivers  to  substitute  a  vacuum- 

Rr.H  tube  voltmeter  for  a  tube  or 

to  place  it  in  parallel  with  a 
tube.  The  inherent  capaci- 
tance of  the  stage  is  usually  so 
low  that  accurate  measure- 
ments cannot  be  made  if  the 
measurement  setup  appreci- 
ably changes  its  value.  A 
preferred  method  is  as  follows 
(Fig.  3) :  A  resistor  of  approxi- 
mately 500  ohms  is  substi- 
tuted for  the  normal  plate 
load  of  the  tube  which  the 
vacuum-tube  voltmeter 
would  replace  at  lower  fre- 
quencies. Under  most  condi- 
tions it  is  inadvisable  to  shunt 
this  resistor  across  the  normal 
plate  load.  The  low-potential 
end  of  the  resistor  is  by- 
passed directly  to  ground, 
keeping  lead  lengths  as  short 
as  possible.  The  signal  gener- 
ator is  then  connected  to  the 
grid  of  the  tube,  the  normal 
grid  load  being  disconnected, 
and  a  vacuum-tube  voltmeter 
is  connected  across  the  resis- 
tor in  the  plate  circuit.  The 
input-output  voltage  charac- 
teristic can  then  be  checked 
and  a  suitable  operating  point 
chosen  on  the  linear  portion 
of  the  gain  characteristic. 
With  the  suggested  value  of 
load  resistance  and  the  usual 
high-transconductance  tube, 
a  gain  of  0-6  db  will  be  se- 
cured. 


B-f 


A.V.6. 


FIG.   3. 

B,  calibrating  rf 


e   Measurement  Set-up.     A,   normal   circuit; 
e;  C,  set-up  for  measurement  of  antenna  gain 
and  bandwidths. 


SINGLE-STAGE  MEASUREMENTS 


11-53 


Under  these  conditions  it  will  be  convenient  to  select  as  a  reference  output  voltage  that 
value  which  corresponds  to  a  signal  on  the  grid  of  10  db  below  1  volt.  It  is  important  that 
the  voltmeter  have  sufficient  sensitivity  to  make  possible  the  selection  of  an  operating 
point  which  can  be  obtained  without  danger  of  grid  current  due  to  high  signal  levels  on 
the  grid.  After  the  operating  point  has  been  chosen,  the  grid  of  the  tube  can  be  connected 
back  in  the  circuit,  the  signal  generator  connected  to  the  grid  of  the  preceding  tube,  and 
performance  measurements  made  without  upsetting  the  grid  loading  in  the  stage  being 
measured. 

ANTENNA  GAIN  AND  BANDWIDTHS.  The  measurement  of  antenna  gain  and 
band  width  to  provide  data  representative  of  those  secured  under  actual  operating  condi- 
tions requires  considerable  care.  If  the  first  tube  is  a  pentode  r-f  amplifier,  antenna  gain 
and  band  widths  may  be  measured  by  placing  the  voltmeter  across  a  resistive  plate  load 
of  a  few  hundred  ohms  and  then  calibrating  the  tube  as  previously  described.  The  data 
may  then  be  secured  without  upsetting  actual  operating  conditions. 

If  the  antenna  secondary  feeds  into  the  grid  of  the  converter  tube,  it  may  not  be  possible 
to  measure  antenna  gain  separately.  In  such  cases,  it  is  usually  possible  to  measure  the 
gain  from  the  antenna  terminals  to  a  resistive  load  in  the  plate  circuit  of  the  i-f  amplifier 
tube.  In  this  way  the  sum  of  the  antenna  gain  and  conversion  gain  can  be  obtained.  If 
it  is  possible  to  measure  the  conversion  gain  separately,  the  antenna  gain  can  be  obtained 
indirectly  by  subtraction  of  the  conversion  gain. 

R-F  GAIN  AND  BANDWIDTH.  The  method  of  measurement  of  r-f  gain,  like  that  of 
the  measurement  of  antenna  gain,  depends  to  a  great  extent  on  the  type  of  circuit.  In 
some  cases,  the  voltmeter  can  be  shunted  across  the  grid  of  the  modulator  tube  to  enable  a 
measurement  of  r-f  gain  to  be  made.  In  others,  it  is  necessary  to  measure  the  r-f  gain  plus 
the  conversion  gain  as  has  been  discussed  under  the  immediately  preceding  heading.  It 
is  always  necessary  to  exercise  care  to  make  sure  that  the  loading  on  the  r-f  tuned  circuit 
under  measurement  conditions  is  the  same  as  that  under  actual  operating  conditions. 

CONVERSION  GAIN.  Conversion  gain  measurements  where  a  pentagrid  converter 
is  used  can  generally  be  made  by  placing  the  vacuum-tube  voltmeter  across  a  resistive 
load  in  the  i-f  plate  circuit  as  has  previously  been  discussed  for  antenna  and  r-f  gain 
measurements.  Where  a  pentode  modulator  is  used  with  both  oscillator  and  signal  voltages 
on  the  same  grid,  it  is  necessary  to  measure  conversion  gain  indirectly.  The  antenna  gain 
plus  conversion  gain  or  r-f  gain  plus  conversion  gain  can  be  measured.  The  antenna  gain 
or  r-f  gain,  as  the  case  may  be,  can  then  be  measured  and  subtracted  to  give  the  conversion 
gain.  As  usual,  in  making  measurements  at  these  frequencies,  care  must  be  exercised  so 
as  not  to  allow  the  measurement  setup  to  change  the  loading  on  the  antenna  coil. 

I-F  GAIN  AND  BANDWIDTH.  The  measurement  of  i-f  gain  and  band  width  at  the 
standard  intermediate  frequency  of  10.7  megacycles  can  best  be  made  by  placing  the 
vacuum-tube  voltmeter  across  a  resistive  load  in  the  plate  circuit  of  the  tube  following 
the  stage  on  wjiich  measurements  are  being  made.  The  tube  can  then  be  calibrated  and 
the  measurements  made  without  upsetting  actual  operating  conditions. 

REJECTION  OF  A-M  BY  F-M  DETECTOR.  No  measurement  has  yet  been  standard- 
ised for  this  test.  However,  Fig.  4  shows  one  method  for  setting  up  equipment  in  a  way 
which  will  provide  useful 
information  regarding  the 
ability  of  an  f-m  detector 
to  reject  amplitude  modu- 
lation. Imperfect  a-m  re- 
jection at  any  signal  input 
level  will  result  in  the 
"bow-tie"  pattern.  For 
this  test  the  f-m  signal 
generator  must  be  capable 
of  being  simultaneously 
amplitude-  and  frequency- 
modulated.  The  a-m  and 
f-m  frequencies  should  be 
asynchronous.  It  is  sug- 
gested that  the  f-m  fre- 
quency be  approximately 
100  cycles  and  that  the 
a-m  frequency  be  approxi- 
mately 400  cycles. 

OSCILLATOR  DRIFT. 
The  frequency  drift  of  an 
f-m  receiver  oscillator  cir- 


FIG.  4.     Equipment  Set-up  for  Test  of  A-m  Rejection  in  F-m  De- 
tectors 


11-54  ELECTRICAL  MEASUREMENTS 

cuit  may  be  determined  by  employing  the  equipment  setup  shown  in  Fig.  5.    The  procedure 
is  as  follows: 

1.  Tune  the  receiver  to  the  100-megacycle  signal  from  the  crystal  oscillator. 

2.  Adjust  the  frequency  and  level  to 
produce  zero  beat  in  the  output. 

3.  Maintain  aero  beat  by  adjusting  the 
frequency  of  the  signal  generator.    Note 
frequency  drift  versus  time  until  the  re- 
ceiver has  stabilized. 

^-Q-7  Mc  MISCELLANEOUS    MEASURE- 

MENTS.    There  are  several  character- 
istics for  which  measurement  procedures 
have  not  yet  been  standardized.     These 
•  AocTto   include  oscillator  radiation,  effect  of  de- 

,  ( ,      I  outpcrf  tuning,  two-signal  interference  tests,  tests 

Fro.  5.    Oscillator  Drift  Measurement  Set-up        °n  built-in  antennas   tests  for  the  effects 

of   downward    modulation,    and    others. 

The  measurement  of  such  characteristics  is  important,  but  a  lack  of  standardized  pro- 
cedures precludes  the  possibility  of  discussion  at  this  time. 

BIBLIOGRAPHY 

I.R.E.  Standards  on  Radio  Receivers,  Latest  issue.    Methods  of  Testing  F-M  Receivers,  1947.    Methods 

of  Testing  A-M  Receivers,  1948. 

Foster,  D.  E.,  Receiver  Characteristics,  Communications,  May  1939. 
Characteristics  of  Broadcast  Receivers,  The  R.M.A.  Engineer,  May  1940. 

Matthews,  A.  C.,  Measurement  of  Receiver  Characteristics,  Radio,  December  1944,  March  1945. 
Smith,  F.  Langford,  Radiotron  Designers'  Handbook,  Third  Edition,  Chapter  29,  pp.  227-239. 
Swinyard,  W.  <X,  Measurement  of  Loop  Antenna  Receivers,  Proc.  I.R.E.,  Vol.  29,  No.  7  (July  1941). 


WAVE  ANALYSIS 

By  E.  Peterson 

Some  of  the  terms  applied  to  devices  which  isolate,  measure,  and  display  the  component 
amplitudes  of  complex  electric  waves  include  wave,  harmonic,  and  spectrum  analyzers; 
distortion  and  intermodulation  meters  and  analyzers;  and  spectrographs. 

28.  WAVE  CHARACTERISTICS 

Waves  to  be  analyzed  may  have  components  located  at  discrete  frequencies  or  dis- 
tributed more  or  less  generally  throughout  frequency  bands.  Discrete  components  are  en- 
countered in  studies  of  the  response  of  a  non-iinear  circuit  or  circuit  element  to  the  action 
of  periodic  waves.  The  resultant  modulation  products  can  be  localized  and  measured 
individually  or,  in  some  cases,  collectively.  Band  distributions  on  the  other  hand  are 
found  in  speech  and  music  and  sometimes  in  noise.  There,  where  the  spectral  distribution 
changes  with  time,  it  is  difficult  to  keep  track  of  individual  components,  and  measurements 
are  conveniently  made  of  energies  within  bands  of  definite  extent. 

DISCRETE  FREQUENCY  DISTRIBUTIONS.  Discrete  or  single  frequency  analysis 
is  used  in  a  variety  of  fields  such  as  noise  reduction,  machinery  noise,  power  interference, 
and  modulation  studies.  Of  these  the  last  mentioned  is  the  most  important  and  the  most 
demanding.  Modulation  studies  are  of  interest  throughout  the  frequency  range  occupied 
by  the  various  communication  services,  and  the  amounts  of  power  associated  with  them, 
vary  greatly  in  the  different  applications.  Considering  extreme  cases,  the  power  associated 
with  modulation  products  in  long  loaded  cables  and  in  multichannel  carrier  amplifiers  may 
be  comparable  with  resistance  noise  in  an  audio  band,  while  modulation  products  in  high- 
power  radio  transmitters  may  amount  to  kilowatts. 

The  frequency  of  any  modulation  product  is  expressible  in  terms  of  the  fundamental 
frequencies  which  produce  it.  If  as  example  there  are  two  fundamental  frequencies  A 
and  /i,  the  frequencies  of  all  products  are  included  within  the  expression  \rnfi  ±  nfz\. 
Here  m  and  n  are  positive  integers,  or  zero.  Products  may  be  distinguished  by  their 
order,  which  is  the  sum  of  m  and  n.  Thus  the  components  2/i,  2/2,  |/i  zh  /a|  are  all  of  the 
second  order;  3/i,  3/2,  \2fi  =b  /2|,  ]/i  =fc  2/2|  are  all  of  the  third  order. 


WAVE   CHARACTERISTICS  11-55 

Modulation  product  amplitudes  may  be  expressed  in  terms  of  the  impressed  waves  or 
fundamentals  producing  the  modulation,  or  in  terms  of  a  significant  component,  using 
arithmetic  or  logarithmic  (decibel)  scales  for  the  ratios.  Whereas  in  many  detailed  studies 
each  modulation  product  of  importance  is  specified  individually,  in  others,  in  which  an 
idea  of  the  total  modulation  is  required,  the  rms  value  of  all  unwanted  products  within  a 
certain  frequency  band  is  specified.  Thus,  in  one  method  of  testing  distortion  in  a  radio 
link  a  sine-wave  signal,  preferably  at  a  low  audio  frequency,  is  impressed  upon  the  trans- 
mitter. At  the  receiver  output  this  component  is  removed  by  a  filter  or  bridge  type  of 
suppression  network.  The  rms  value  of  all  remaining  products  in  the  audio  band  is 
measured  by  means  of  a  thermocouple  or  suitable  rectifier.  When  expressed  as  a  per- 
centage of  the  rms  fundamental  in  the  normally  terminated  circuit,  the  ratio  is  termed  the 
distortion  factor. 

When  the  modulation  of  a  particular  circuit  element  is  under  consideration,  and  is  to 
be  expressed  in  a  way  which  would  be  applicable  with  the  element  connected  in  any  given 
circuit,  it  is  customary  to  specify  the  equivalent  (modulation)  generator  emf.  This  is 
the  voltage  which,  when  inserted  in  the  circuit  made  up  of  linear  elements,  gives  rise  to 
the  observed  distortion  products.  It  is  used  ordinarily  of  elements  only  slightly  non-linear. 

The  ratios  of  modulation  products  to  their  fundamentals  which  analyzers'  are  used  to 
evaluate  fall  into  two  ranges  of  values.  One  applies  to  products  falling  within  the  frequency 
range  constituting  quality  impairment  of  the  originating  channel.  The  other  characterizes 
products  which  fall  outside  the  transmitted  frequency  range,  constituting  interference 
with  other  channels.  Examples  may  be  drawn  from  the  audio,  carrier,  and  radio  broadcast 
fields.  Thus  the  requirements  on  a  system  transmitting  speech  are  usually  that  the  dis- 
tortion products  be  at  least  20  to  40  db  down  on  the  wanted  components  in  the  normal 
channel  ( 10  to  1  per  cent  in  amplitude,  or  1  to  0.01  per  cent  in  power) .  Interfering  products 
in  other  channels  are  restricted  to  smaller  amplitudes.  In  multichannel  carrier  telephone 
amplifiers  the  figures  range  from  40  to  80  db  down ;  in  multichannel  carrier  grouping  filters 
which  serve  to  separate  transmitted  and  received  currents  the  figures  range  from  70  to 
120  db  down.  In  certain  radio  transmitters  the  products  appearing  in  other  channels  are 
restricted  to  something  of  the  order  of  70  db  down  on  the  fundamentals.  Corresponding 
figures  for  elements  or  units  which  make  up  a  system  may  be  more  stringent  than  those 
specified  for  the  system  as  a  whole,  especially  where  the  modulations  in  several  elements 
or  units  contribute  to  the  total. 

TEST  WAVES.  In  the  analysis  of  speech,  music,  or  noise,  or  of  the  output  waves  of 
such  devices  as  oscillators  and  harmonic  producers,  the  questions  involved  are  those  of 
isolating  and  measuring  the  components.  In  establishing  the  performance  of  circuits  or 
of  circuit  elements,  however,  the  additional  problem  arises  of  specifying  and  of  supplying 
the  wave  to  be  used  for  test  purposes. 

Generally  speaking,  the  input  wave  should  permit  an  indication  of  the  performance 
of  the  apparatus  to  be  tested,  approximating  as  closely  as  possible  the  conditions  of  actual 
use.  Where  the  apparatus  is  subjected  in  use  to  an  impressed  wave  of  simple  form,  com- 
paratively little  difficulty  arises  since  the  wave  can  be  readily  reproduced  by  oscillators 
and  filter  networks.  On  the  other  hand,  the  apparatus  may  be  subjected  to  signal  waves 
which  are  highly  complex  in  nature,  such  as  those  of  speech.  Such  waves  may  require 
comparatively  elaborate  and  cumbersome  setups  for  the  measurement  of  modulation 
products,  especially  where  the  fundamentals  and  the  various  products  overlap  in  the  fre- 
quency spectrum. 

To  facilitate  analysis,  the  complex  signal  waves  are  replaced  by  comparatively  simple 
ones.  At  the  present  time  one  of  two  types  is  ordinarily  used  according  to  the  problem 
encountered — a  pure  sine  wave,  or  a  complex  wave  consisting  of  two  sine- wave  components. 
To  prevent  interaction  of  the  two  components  before  reaching  the  circuit  under  test,  tuned 
circuits,  filter  networks,  or  bridge  networks  may  be  employed.  The  amplitude  of  the 
testing  wave  is  chosen  so  as  to  traverse  much  the  same  region  of  the  non-linear  character- 
istic as  does  the  complex  signal  wave  for  which  it  is  substituted.  In  some  cases  the  test 
wave  is  made  to  have  the  same  rms  value,  and  in  others  the  same  peak  value,  as  the  normal 
signal.  The  test  frequencies  are  so  selected  that,  together  with  their  important  modulation 
products,  they  fall  in  a  frequency  region  of  interest. 

To  illustrate  the  use  of  single-  and  two-frequency  test  waves  consider  the  investigation 
of  a  narrow  band  amplifier  passing  frequencies,  let  us  say,  from  100  to  110  kc.  If  a  sine 
wave  of  frequency  105  kc  is  impressed  for  test  purposes  then  the  harmonics  are  210  kc, 
315  kc,  and  so  on,  all  harmonics  falling  outside  the  transmitted  band  and  being  greatly 
attenuated.  A  measurement  of  distortion  within  the  pass  band  would  yield  nothing.  If, 
on  the  other  hand,  instead  of  applying  a  single  frequency,  we  impressed  two  frequencies 
at  105  and  106  kc,  say,  the  harmonics  would  fall  outside  the  band  and  be  attenuated  as 
before,  but  the  third-order  products  at  104  and  107  kc  would  lie  within  the  band,  as  indi- 


11-56 


ELECTRICAL  MEASUREMENTS 


cated  in  the  spectrum  (Fig.  1).  These  would  furnish  a  useful  indication  of  the  non-linear 
distortion  which  a  single-frequency  test  wave  could  not  provide.  Another  example  may 
be  drawn  from  the  same  vacuum-tube  amplifier.  It  may  be  shown  that  the  amplitudes 
of  the  above-mentioned  third-order  products  depend  upon  the  impedance  offered  to  second 
orders,  one  of  which  is  the  difference  frequency  (1  kc  in  the  above  example).  This  imped- 
ance may  vary  widely  in  different  designs  without  affecting  the  normal  transmission  or 


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Frequency  KG 

FIG.  1.     Output  Wave  Spectrum  of  Narrow-band  Amplifier  with  Frequencies  of   105  and  106  kc 

Impressed 

harmonic  production.  Figure  2  shows  the  results  of  measurement  of  third-order  modula- 
tion as  a  function  of  the  frequency  difference  between  the  two  input  components.  No  indi- 
cation of  this  significant  effect  would  be  revealed  by  a  single-frequency  testing  wave. 

Another  instance  in  which  the  two-frequency  test  wave  is  used  is  in  the  measurement 
of  distortion  produced  by  ferromagnetic-core  coils.  There  only  the  main  hysteresis  loops 
are  called  into  play  when  a  sinusoidal  magnetizing  force  is  applied,  but  with  complex 
waves  of  magnetizing  force  a  different  characteristic  is  involved  since  subsidiary  loops 
appear. 

To  exemplify  the  use  of  the  single-frequency  test  wave,  consider  a  vacuum-tube  amplifier 
supplied  with  power  obtained  from  the  60-cycle  line.  There  exists  a  certain  amount  of 
modulation  of  each  signal  component  with  the  60  cycles  and  its  harmonics  in  the  amplifier. 
To  measure  this  effect  a  single-frequency  test  wave  will  serve. 

BAND  FREQUENCY  DISTRIBUTIONS.  Band  frequency  analysis  involves  measure- 
ment of  energies  or  amplitudes  in  the  sub-bands  into  which  the  main  band  is  divided.  It 
is  usually  employed  for  waves  such  as  room  noise  or  windage  noise  from  certain  types  of 
_^  machinery,  which  contain  no  promi- 

nent discrete  frequency  components, 
and  for  waves  which  contain  discrete 
frequency  components  varying  rapidly 
in  both  frequency  and  amplitude,  such 
as  speech  and  music.  Band  frequency 
analyses  have  also  been  used  for  waves 
consisting  largely  of  discrete  frequency 
components  when  it  is  unnecessary  to 
know  the  precise  frequencies  of  the  in- 
dividual components.  Examples  come 
up  in  studying  the  effect  of  vibration- 
absorbing  mountings  and  of  the  acous- 
tical treatment  of  rooms  for  the  reduc- 
tion  of  machinery  noise.  In  the  com- 
FIG.  2.  Third  Order  Modulation  at  Amplifier  Output  munications  field  band  frequency  an- 
Function  of  ^  the  Two  program  materfal  and  of 


Third  Order  db  Down  on  Fundamenla 

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/ 

)                   50                100              150               200 
Difference  Frequency 

circuit  noise  are  useful  in  determining 
requirements  to  be  placed  on  systems  for  reproduction  of  the  input  waves  with  the 
desired  fidelity  and  freedom  from  interference.  For  testing  multichannel  carrier  systems, 
the  loading  effect  of  many  talkers  has  been  simulated  by  introducing  bands  of  resistance 


29.  GENERAL  ANALYZER  REQUIREMENTS 

The  requirements  which  an  analyzer  is  called  upon  to  satisfy  in  general  are  those  ordi- 
narily imposed  upon  any  piece  of  measuring  apparatus :  that  it  furnish  an  indication  of  the 
quantity  sought,  within  certain  limits  of  accuracy  and  of  precision,  without  disturbing  the 
essential  performance  of  the  circuit  to  which  it  is  applied. 


GENERAL  ANALYZER  REQUIREMENTS 


11-57 


INPUT  COUPLING.  Connection  of  the  analyzer  to  the  circuit  under  test  without 
altering  the  essential  circuit  performance  is  accomplished  by  assigning  a  suitable  impedance 
to  the  input  circuit  of  the  analyzer.  Three  circuit  arrangements  have  been  used  for  this 
purpose.  These  are  the  high-impedance  or  voltage-analyzer  connection,  the  low-impedance 
or  current-analyzer  connection,  and  the  termination.  In  the  first  the  analyzer  is  made  to 
have  an  impedance  much  higher  than  that  of  the  circuit  across  which  it  is  shunted.  In 
the  second,  the  analyzer  is  made  to  have  an  impedance  much  smaller  than  that  of  the 
circuit  in  series  with  which  it  is  connected.  In  the  third,  the  analyzer  is  made  to  have  a 
definite  resistance  to  replace  a  resistance  of  the  same  value  in  the  circuit  studied.  It  will 
be  observed  that,  under  these  conditions,  insertion  of  the  analyzer  will  normally  have 
but  little  influence  upon  the  normal  functioning  of  the  circuit  studied,  since  all  three  may 
be  made  balanced  or  unbalanced  to  ground  as  desired.  Figure  3  shows  connections  to  a 
resonance  type  of  analyzer  as  an  example. 
For  current  analysis,  connection  is  made  to 
point  a  and  for  voltage  analysis  to  point  6 
when  the  resistor  R  is  made  large.  For  use  as 
a  termination  R  is  given  a  suitable  fixed 
value  to  fit  the  circuit  or  line  to  which  it  is 
connected,  usually  600  or  72  ohms.  The  re- 
sistance r  is  usually  made  less  than  the  effec- 
tive resistance  of  the  tuned  circuit  to  avoid 
reducing  selectivity  and  may  be  adjusted 
for  the  required  gain. 

A  preferable  connection  for  voltage  analy- 
sis, shown  in  Fig.  3b,  usually  permits  a 
greater  fraction  of  the  desired  input  voltage 
component  to  be  transmitted  while  main- 
taining a  high  input  impedance.  Another 
connection  suitable  for  voltage  analysis, 
particularly  useful  over  wide  frequency 
bands,  employs  a  probe  including  a  con- 
denser-resistance divider  (Fig.  3c).  Its  in- 
put impedance  is  that  of  a  1-  or  2-ju/if  ca- 
pacitance shunted  by  a  resistance  of  the 
order  of  1  megohm.  To  make  transmission  flat  with  frequency,  the  two  time  constants 
C\Ri  and  CzRz  must  be  made  equal. 

RESOLUTION  OF  COMPONENTS.  The  analyzer  band  should  be  wide  enough  to 
permit  easy  tuning-in  of  the  desired  component,  including  an  allowance  for  variations  in 
frequency  of  the  selected  component.  At  the  same  time  it  must  be  narrow  enough  to 
select  a  specific  product.  The  equivalent  band  width  of  a  selective  circuit  may  be  defined 
as  the  band  width  of  an  idealized  filter  having  a  constant  loss  in  the  pass  band  equal  to 
the  minimum  loss  of  the  circuit  in  question,  infinite  loss  outside  the  pass  band,  and  passing 
the  same  amount  of  energy  from  a  continuous  constant-energy  spectrum  applied  to  the 
input.  Analyzers  for  the  study  of  low-frequency  machine  noise  require  a  band  width  from 
3  to  5  cycles.  Those  for  telephone  circuit  noise  caused  by  induction  from  neighboring 
power  circuits  require  an  equivalent  band  width  of  20  to  30  cycles  at  least.  For  speech 
and  for  general  types  of  noise,  band  widths  of  45  to  300  cycles  are  appropriate.  For  inter- 
modulation  studies  of  discrete  components,  the  bands  used  range  from  a  cycle  or  so  at  low 
frequencies  to  several  kilocycles  and  more  at  high  frequencies. 

Similarly  the  frequency  discriminations  required  vary  widely  in  different  cases.  In 
fixed-band  analyzers  used  for  speech,  discriminations  of  20  db  or  so  against  components 
located  at  the  centers  of  neighboring  bands  are  sufficient  to  provide  useful  information. 
Other  a-f  analyzers  usually  require  discriminations  of  the  order  of  25  to  50  db  against 
components  60  cycles  from  the  tuned  frequency. 

The  wide  divergence  in  requirements  has  led  to  the  development  of  a  number  of  different 
forms  of  analyzers,  each  suited  to  a  limited  class  of  work. 

MEASUREMENT  AND  DISPLAY  OF  SELECTED  COMPONENTS.  With  the 
simpler  manually  operated  forms  of  frequency  analyzers,  the  point-by-point  method  of 
analysis  is  employed.  Here,  for  the  discrete  frequency  analysis  of  a  steady  wave,  the 
frequency  spectrum  is  explored  point  by  point  over  the  range  of  interest,  and  whenever  a 
component  of  importance  is  located  its  frequency  and  amplitude  are  observed.  For  band 
frequency  analysis  of  a  periodic  wave,  measurements  of  the  energy  in  each  of  a  number  of 
adjacent  sub-bands  are  made  successively  without  reference  to  the  distribution  within  the 
sub-band.  In  analyzing  waves  of  short  duration  which  can  be  repeated  a  large  number  of 
tunes  during  the  exploration  of  the  frequency  spectrum,  a  similar  procedure  is  followed. 


FIG.  3,     Analyzer  Input  Circuits  Illustrating  High- 
input  Impedance  and  Low-input  Impedance  Con- 
nections 


11-58  ELECTRICAL  MEASUREMENTS 

The  net  result  of  an  analysis  in  which  the  several  components  are  examined  individually 
is  usually  presented  as  a  meter  deflection.  To  permit  the  use  of  simple  d-c  meters  the 
selected  wave  feeds  a  thermocouple  or  a  rectifier  of  the  vacuum-tube  or  copper  oxide  type. 
The  thermocouple  is  somewhat  sluggish  in  operation,  and  the  process  of  tuning  accordingly 
must  be  slowed  sufficiently  to  permit  the  products  to  produce  an  observable  deflection. 
The  slow  response  is  advantageous  where  the  smoothing  out  of  rapid  fluctuations  in  the 
selected  wave  is  desired.  Deflections  are  closely  proportional  to  the  square  of  the  heater 
current.  This  is  a  valuable  property  in  estimating  energies  in  band  analysis.  Tube  and 
varistor  circuits  may  "be  made  much  more  rapid  in  operation  and  may  be  given  a  wide 
range  of  desired  relations  between  input  amplitude  and  deflection  by  suitable  choice^  of 
operating  potentials  and  circuit  parameters.  For  high  rectifying  gain  a  small  negative 
bias  may  be  used.  For  greater  precision  a  large  negative  bias  may  be  used  since  then  a 
small  input  change  may  be  made  to  yield  a  large  change  in  rectified  output.  By  the  same 
token,  however,  an  interfering  component  produces  a  correspondingly  large  change  in 
rectified  output. 

Where  a  graphical  record  of  wave  analysis  is  to  be  had  in  a  short  period  of  time,  one  of 
several  types  of  recording  frequency  analyzers  is  employed.  These  recording  analyzers 
are  designed  to  perform  automatically  the  same  operations  performed  manually  with  the 
simpler  devices,  including  the  plotting  of  wave  component  values.  Recording  analyzers 
of  this  type  may  be  used  as  well  for  the  analysis  of  non-periodic  waves  of  short  duration 
which  san  be  repeated  so  that  the  effect  of  a  steady  wave  is  obtained. 

Thus  in  the  analysis  of  speech  sounds  and  of  single  tones  from  musical  instruments,  or 
of  other  non-steady  waves  which  are  not  readily  reproducible,  with  devices  employing  the 
point-by-point  method,  the  wave  may  be  recorded  and  repeatedly  reproduced.  Finally, 
non-steady  waves  may  be  analyzed  directly  without  the  necessity  of  recording  them  by 
means  of  devices  giving  a  visual  indication  of  the  complete  energy  spectrum  of  the  wave 
and  capable  of  showing  the  rapid  changes  in  spectrum  which  are  characteristic  of  speech 
sounds  and  some  musical  tones.  A  photograph  of  cathode-ray  patterns  or  a  recording  on 
electrically  sensitive  paper  of  the  indications  of  such  an  analyzer  can  be  made  for  a  per- 
manent record  of  the  analysis.  These  arrangements  are  discussed  in  article  31,  p.  11-65. 

30.  METHODS  OF  WAVE  ANALYSIS 

The  earliest  methods  used  for  wave  analysis  were  based  upon  observed  wave  forms — 
observed  directly  by  means  of  an  oscillograph,  or  synthesized  point  by  point  from  syn- 
chronous contactor  measurements.  Wave  components  are  deducible  from  the  wave  form 
by  application  of  Fourier  series,  which  expresses  the  amplitude  of  any  component  as  an 
integral.  Evaluation  of  the  integrals  is  customarily  handled  by  a  time-consuming  point- 
by-point  calculation,  for  which  tables  and  schedules  have  been  formulated  to  facilitate  the 
work.  This  method  is  useful  when  only  harmonics  of  a  single  frequency  are  present  in  the 
wave  to  be  analyzed;  when  the  wave  has  more  than  one  fundamental,  the  computation 
ordinarily  becomes  too  involved  to  be  useful,  since  a  multiple  Fourier  series  must  be  em- 
ployed. For  harmonics  of  a  single  frequency  which  are  not  too  small  in  amplitude,  the 
accuracy  of  the  determination  depends  largely  upon  the  accuracy  with  which  the  oscillo- 
gram  is  read,  the  number  of  points  included  in  the  analysis,  and  the  order  of  the  harmonic. 
Because  of  its  limitations,  the  computational  method  has  been  superseded  by  methods  of 
direct  measurement,  wherever  possible. 

The  first  direct  method  of  analysis  was  that  of  Pupin,  in  which  a  series-resonant  circuit 
was  used  to  select  a  component  of  interest,  and  amplitude  was  deduced  from,  the  indication 
of  an  electrostatic  voltmeter  connected  across  the  condenser  of  the  tuned  circuit.  Discrim- 
ination by  means  of  simple  tuned  circuits  is  employed  in  certain  types  of  analyzers  of  the 
present  day,  their  sensitivity  and  flexibility  being  greatly  increased  by  means  of  vacuum- 
tube  amplifiers.  Selectivity  and  sensitivity  are  improved  by  having  several  tuned  circuit 
and  amplifier  units  in  cascade,  and  by  substituting  filter  networks  for  the  simple  resonant 
circuit.  A  different  way  to  improve  selectivity  involves  the  use  of  a  negative  resistance 
circuit  to  reduce  tuned  circuit  losses. 

RESONANCE  ANALYZERS.  Variable  Timed  Circuit.  The  simple  form  of  current 
analyzer  shown  in  Fig.  4  is  useful  where  a  high  degree  of  frequency  selectivity  is  not  re- 
quired. It  is  made  up  of  four  units:  coupling  unit,  frequency-discriminating  network, 
amplifier,  and  indicator.  The  tuning  unit  precedes  the  amplifier  to  discriminate  against 
other  components  of  the  input  wave.  This  procedure  reduces  modulation  in  the  amplifier, 
which  may  introduce  components  of  the  same  frequencies  as  those  to  be  measured. 

The  tuning  unit  consists  of  a  simple  series-resonant  circuit  with  the  input  wave  intro- 
duced through  a  small  resistor.  The  amplifier  input  is  taken  across  the  condenser  when 


METHODS  OF  WAVE  ANALYSIS 


11-59 


FIG.  4.     Resonance  Analyzer  Circuit   with  Input 
Arranged  for  Current  Analysis t 


the  input  fundamentals  are  of  frequencies  higher  than  the  frequency  of  the  component 
being  measured.  When  the  fundamentals  or  other  possible  interfering  components  are  at 
Lower  frequencies  the  amplifier  input  is  connected  across  the  coil  for  greater  discrimination. 
Ine  amplifier  is  made  responsive  over  the 
frequency  range  of  interest;  it  is  usually  of 
the  resistance-capacitance  coupled  type. 

To  increase  the  selectivity  over  that 
available  in  the  simple  tuned  circuit,  several 
discrirninator-amplifier  units  are  used  in 
cascade.  Efficient  coupling  between  units 
is  provided  by  step-down  transformers. 
Here  the  signal  level  must  be  kept  low- 
enough  to  avoid  appreciable  modulation  of 
the  fundamentals  in  the  early  amplifier 
stages.  A  preferable  arrangement  from  this 
standpoint  would  be  to  locate  as  much  of 
the  necessary  frequency  discrimination  as 

possible  ahead  of  the  amplifiers.  For  greater  discrimination  than  simple  tuned  circuits 
can  conveniently  afford,  use  is  made  of  filter  networks,  which,  being  much  more  cumber- 
some and  less  flexible  in  adjustment,  are  usually  employed  where  only  a  small  number  of 
fixed  frequencies  are  to  be  measured. 

The  substitution  method  is  currently  used  for  evaluating  component  amplitudes  since 

the  loss  of  the  tuned  circuit,  and  therefore 
the  analyzer  response,  varies  with  fre- 
quency. The  connections  are  shown  in 
Fig.  5,  the  switch  being  used  to  connect  the 
analyzer  to  either  the  test  or  the  standard 
circuits.  The  standard  circuit  includes  an 
oscillator  adjustable  in  frequency,  a  vao- 

FIG.O.    Substitution-Method  for  Evaluating  Com-    umn-tube  voltmeter  or  thermocouple  and 
ponent  Amplitudes  associated  meter  to  indicate  the  standard 

input,  and  a  calibrated  attenuator  to  vary 

the  input  amplitude  to  the  analyzer.  The  analyzer  input  impedance  is  fised  at  a  value 
at  which  the  attenuator  is  properly  terminated. 

The  test  procedure  consists  in  connecting  the  analyzer  to  the  unknown,  tuning  in  the 
desired  product,  and  adjusting  the  analyzer  gain  until  a  suitable  deflection  is  obtained  on 
the  output  meter.  The  switch  is  then  thrown  to  the  standard  side,  and  the  standard 
oscillator  frequency  is 
varied  until  it  coincides 
with  the  analyzer  tuning. 
The  attenuator  is  then 
varied  until  the  same  de- 
flection is  obtained  as  be- 
fore. The  frequency  of 
the  unknown  component 
may  then  be  found  from 
the  frequency  calibration 
of  the  standard  oscillator 
or  of  the  analyzer,  and 
the  amplitude  is  com- 
puted from  the  voltmeter 
reading  and  attenuator 
setting. 

Another  form  of  reso- 
nance analyzer  with  shunt 
tuning  in  the  plate  circuits 
of  two  amplifier 


Ivolts  across  600  Ohris  M  ,_,  ,_ 
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1  1 

i 

3      400     800 

1200    1600    2000    2400    2800 

FIG.  6. 


Frequency:  Cycles  per  Second 
Spectrum  of  Noise  on  an  Open-wire  Telephone  Circuit 


has  been  used  extensively 
in  the  analysis  of  power- 
frequency  harmonics  and 
of  induction  from  power 
circuits.  Voltages  from  5  X  10  ~6  to  50  are  measurable  in  the  range  up  to  3  kc  with  an 
accuracy  of  d=5  per  cent.  The  discrimination  of  such  an  analyzer  varies  with  frequency, 
being  about  40  db  at  60  cycles  from  the  tuned  frequency  when  tuned  to  180  cycles,  and  from 
24  to  32  db  60  cycles  away  when  tuned  to  3000  cycles.  Figure  6  gives  an  example  of  an 


11-60 


ELECTRICAL  MEASUREMENTS 


analysis  of  telephone-circuit  noise  caused  by  induction  from  power  circuits.  Approximately 
1  hour  is  required  to  obtain  with  this  apparatus  the  amount  of  data  shown  in  the  ngure. 

The  speed  and  ease  of  operation  of  the  more  modern  analyzers  have  resulted  in  supplant- 
ing the  resonance  type  for  all  but  special  purposes. 

Fixed  Bands.  A  decidedly  different  form  of  analyzer  using  direct  frequency  selection 
avoids  the  need  for  manual  tuning  to  each  frequency  of  interest  by  means  of  a  bank  of 
fixed  contiguous  band-pass  niters.  Fourteen  filters  are  used  in  one  particular  model  to 


Entire  Spectrum 

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Frequency  in  Cycles  per  Second 

FIG.  7.     Average  Speech  Pressures  per  Frequency  Interval  of  1  cycle  per  second — normal  conversational 

voice.     Distance  2". 

cover  the  frequency  range  from  SO  to  12,000  cycles  in  approximately  logarithmic  steps. 
The  filters  are  connected  in  parallel  at  the  input  side  and  introduce  a  loss  of  about  8  db  at 
the  crossover  points  and  greater  than  70  db  over  most  of  the  frequency  range  beyond  the 
transmission  band,  A  flux  meter  is  used  to  integrate  the  sub-band  output  over  a  15-second 
interval  for  the  measurement  of  average  amplitudes,  while  a  series  of  gas-filled  discharge 
tubes  with  associated  relays  and  electrical  counters  measures  peak  magnitudes  over  a 
54-db  range  in  steps  of  6  db.  These  measurements  are  made  in  one  band  at  a  time  although 
both  average  and  peak  measurements  of  the  total  spectrum  may  be  made  simultaneously 
with  the  band  measurements.  Peak  measurements  in  Vs-second  intervals  are  made  over 
as  long  a  time  as  desired.  Figure  7  shows  amplitude-frequency  distributions  for  con- 
versational speech  for  male  and  female  voices. 

Another  type  of  fixed  band  analyzer  has  been  used  to  measure  distortion  generated  in 
non-linear  circuits  by  complex  input  waves.  Here  the  input  wave  is  made  to  pass  through 
a  narrow  band-elirnination  filter  before  it  enters  the  circuit  to  be  tested.  A  narrower  band- 
pass filter  is  used  for  the  analysis,  with  its  pass  band  located  within  the  band  eliminated 
at  the  input.  By  this  procedure  the  band-pass  filter  output  has  had  the  fundamental 
components  eliminated,  so  that  it  constitutes  a  measure  of  distortion. 

Feedback  Type.  A  third  means  of  providing  frequency  discrimination  is  particularly 
useful  for  the  analysis  of  discrete  components  in  sub-audio-  and  audio-frequency  ranges. 

It  consists  of  a  negative  feedback  amplifier 
TO  incorporating  a  bridge  type  of  network  in 
RECTIFIER  its  feedback  path.  One  form  shown  in  Fig. 
8  uses  a  parallel  T  network  made  up  of 
capacitances  and  resistances  which  provides 
high  suppression  at  a  single  frequency.  At 
this  frequency  therefore  the  full  amplifier 
gain  is  effective.  At  frequencies  removed 
from  the  balance  point,  transmission 
through  the  feedback  network  acts  degener- 
atively  to  reduce  the  amplifier  gain.  By 
insertion  in  the  feedback  path,  therefore, 
the  null  of  the  network  is  transformed  into  a  transmission  maximum  through  the  amplifier 
circuit.  The  network  null  may  be  conveniently  varied  in  frequency  by  ganging  the  three 
resistors.  Its  selectivity  remains  constant  on  a  percentage  basis  throughout  the  band, 
independent  of  frequency. 


Discrimination 


METHODS  OF  WAVE  ANALYSIS 


11-61 


The  following  figures  on  frequency  discrimination  apply  to  a  two-stage  amplifier  with  a 
null  set  at  25  cycles.  Roughly  12  per  cent  away  from  the  null,  the  discrimination  is  10  db; 
20  per  cent  away  it  is  16  db,  and  in  the  limit  approaches  45  db  down  on  the  maximum 
transmission. 

Transmission  of  the  amplifier  proper  is  made  substantially  fiat  over  the  maximum  range 
of  analysis.  Beyond  that,  the  gain  and  phase  around  the  feedback  loop  are  arranged  to 
avoid  regeneration  and  oscillation,  as  in  feedback  amplifiers  generally. 

SUPPRESSION  AND  INTERMODTTLATION  ANALYZERS.  Although  these  two 
analyzer  forms  are  not  related  in  general,  it  will  be  convenient  here  to  take  them  together. 

A  simple  type  of  suppression  analyzer  previously  mentioned  evaluates  the  rms  sum  of 
the  harmonics  generated  with  usually  a  sinusoidal  signal  impressed.  The  fundamental 
may  be  suppressed  through  the  action  of  a  resonant  bridge,  or  a  suppression  filter,  or  a 
twin-T  network.  The  output  includes  any  noise  and  interfering  components  which  may 
be  present.  If  the  harmonics  are  comparable  in  amplitude  to  these  components,  the 
harmonic  distortion  proper  cannot  be  found  directly  by  this  method. 

Another  form  of  suppression  analyzer  uses  special  procedures  which  are  applicable  to 
the  detection  of  a  carrier  and  its  two  sidebands,  as  practiced  in  radio  broadcast  reception. 
These  procedures  form  the  basis  for  a  method  of  measuring  intermodulation  between  two 
frequencies  in  the  audio  band,  one  low  (40  to  100~)  and  the  other  comparatively  high 
(1  to  12  kc) .  The  two  tones  are  supplied  to  the  system  under  test,  and  the  output  is  filtered 
to  suppress  the  low-frequency  tone.  The  residual  wave  is  then  treated  as  a  carrier  accom- 
panied by  the  two  sidebands  to  determine  the  extent  to  which  the  higher-frequency  com- 
ponent is  modulated  by  the  lower,  as  described  below. 

Measurement  of  Percentage  Modulation.  The  special  procedures  referred  to  may 
involve  the  cathode-ray  oscillograph  or  the  linear  rectifier.  If  the  modulating  signal  is 
applied  to  the  horizontal  deflection  plates  of  the  oscillograph,  and  the  r-f  output  consisting 
of  the  carrier  and  its  two 

second-order   side  frequen- _^ ^       JL- 

cies  is  applied  to  the  vertical  ^ ^  ----- 

deflection  plates,  then  pat- 
terns  like  those  shown  in 
Figs.  9a  and  95  are  obtained 
on  the  oscillograph  screen, 
according  to  the  phase  shifts 
in  the  system.  These  pat- 
terns yield  the  percentage 
modulation  from  the  lengths  of  the  minimum  and  maximum  ordinates  indicated.  One 
hundred  per  cent  modulation  is  obtained  when  the  wave  envelopes  go  to  zero,  and  greater 
modulations  are  indicated  by  intersection  of  the  envelopes.  When  a  linear  sweep  circuit 
controlled  by  the  modulating  signal  is  connected  to  the  horizontal  deflection  plates,  rather 
than  the  modulating  signal  itself,  then  Fig.  9c  is  obtained. 

Knowledge  of  the  percentage  modulation  permits  the  second-order  sideband  amplitude 
(KP/2,  in  the  figure)  to  be  calculated  accurately  only  when  higher-order  sidebands  of  the 
carrier  are  negligibly  small.  In  that  event  the  envelopes  of  Figs.  9a,  97>,  and  9c  are  respec- 
tively linear,  ellipsoidal,  and  sinusoidal.  Otherwise  the  wave  envelopes  may  be  analyzed 
to  yield  the  amplitudes  of  the  second-  and  higher-order  sidebands. 

Another  method  for  determining  percentage  modulation,  generally  used  to  evaluate 
intermodulation,  employs  a  linear  rectifier  (envelope  detector)  to  detect  the  modulated 
carrier  wave  with  a  minimum  of  audio  distortion.  The  percentage  modulation  is  then  given 
as  100  times  the  ratio  of  the  peak  audio  signal  output  to  the  d-c  component.  Where 
higher-order  products  are  present,  they  may  be  evaluated  by  resonance  or  heterodyne- 
type  analyzers. 

HETERODYNE  ANALYZERS.  Dynamometer.  Another  method  of  analysis  first 
used  by  Descoudres  employs  a  dynamometer,  in  which  the  unknown  wave  and  a  standard 
wave  of  known  frequency,  amplitude,  and  phase  are  respectively  connected  to  the  two 
coils  of  the  dynamometer.  Since  the  dynamometer  deflection  is  proportional  to  the  product 
of  the  currents  in  the  two  coils,  a  constant  deflection  is  obtained  when  the  frequency  of 
the  sinusoidal  standard  is  made  equal  to  that  of  a  component  of  the  unknown  wave.  The 
magnitude  and  phase  of  the  component  are  then  found  from  a  calibration  when  the  stand- 
ard phase  is  adjusted  for  maximum  deflection.  To  determine  the  magnitude  without 
regard  to  phase,  the  standard  frequency  is  offset  by  a  fraction  of  a  cycle  from  that  of  the 
component  under  measurement,  and  the  maximum  swing  is  observed.  The  method  is 
limited  to  comparatively  low-frequency  work. 

From  one  standpoint  the  dynamometer  may  be  regarded  as  fulfilling  three  functions: 
modulating  the  unknown  with  the  standard  wave,  filtering  the  beat  frequency  output,  and 


(a) 


Time 

a?)  ro 

FIG.  9.     Modulated  Wave  Patterns 


11-62 


ELECTRICAL  MEASUREMENTS 


indicating  the  amplitude  of  the  difference  frequency  component.  Improvement  in  fre- 
quency response  and  in  sensitivity  has  been  obtained  by  replacing  the  dynamometer  by  a 
vacuum-tube  modulator,  and  further  increase  in  sensitivity  is  obtained  again  by  amplifying 
the  beat  frequency  output.  As  an  indication  of  the  component  to  be  measured,  use  may 
be  made  of  either  the  lower  sideband  or  the  upper  sideband  formed  by  the  beating  oscillator 
and  the  component  under  investigation.  To  select  this  product,  mechanical,  electrical,  or 
piezoelectric  niters  and  networks  are  conveniently  located  in  the  frequency  range.  These 
replacements  of  the  functions  discharged  by  the  dynamometer  result  in  the  most  widely 
used  of  all  analyzers,  the  modern  heterodyne  type. 

A  particularly  simple  form  of  heterodyne  analyzer  has  been  adapted  to  the  measurement 
of  products  not  too  small  in  relative  and  absolute  amplitudes.  In  this  arrangement  no 
amplifier  is  used,  and  the  mechanical  movement  of  a  meter  connected  in  the  plate  circuit 
of  a  modulator  tube  serves  to  provide  frequency  discrimination  as  it  does  in  the  dynamom- 
eter. 

General  Tube  Modulator  Type.  The  heterodyne  analyzer  possesses  a  number  of  advan- 
tages over  other  types  which  have  brought  it  into  wide  use.  To  mention  the  outstanding 
ones,  first  of  all  it  uses  a  fixed  discriminating  circuit  which  is  readily  made  highly  selective 
and  stable,  and  takes  up  little  space-  Next,  the  tuning-in  of  a  desired  band  requires  but  a 
single  adjustment — the  frequency  of  the  heterodyning  oscillator.  Finally  the  response  can 
be  made  fiat  over  an  extended  frequency  range  so  that  the  substitution  method  of  evaluat- 
ing amplitudes  is  not  required.  Relative  levels  can  be  taken  directly  as  the  difference 
between  attenuator  settings. 

Essential  units  of  a  representative  model  are  shown  in  the  schematic  of  Fig.  10.  The 
principal  elements  comprise  coupling  unit,  modulator,  beating  oscillator,  filter,  amplifier, 
and  indicator. 


COUPLING         HETERODYNE       MODULATOR        SELECTIVE 
OSCILLATOR  NETWORK 

FIG.  10.     Heterodyne  Analyzer  Circuit 

The  coupling  unit  is  shown  unbalanced  to  ground  with  a  calibrated  attenuator  which 
serves  to  vary  the  input  to  the  modulator.  The  analyzer  therefore  is  used  as  a  termination, 
but  other  coupling  units  for  voltage  or  current  analysis  can  be  fitted  to  the  input  terminals 
of  the  attenuator. 

The  modulator  is  of  the  conjugate  input  type,  to  provide  a  convenient  means  for  sepa- 
rating the  signal  and  beat  frequency  circuits,  to  balance  out  the  beat  frequency  oscillator 
wave  in  the  output  circuit,  and  to  reduce  the  number  of  unwanted  modulation  components 
produced.  Triodes  have  been  used  with  the  grid  bias  set  close  to  plate-current  cut-off. 
The  shielded  and  balanced  input  coil  shown  should  have  uniform  transmission  over  the 
frequency  range  of  interest,  and  its  modulation  should  be  well  below  the  amplitudes  of 
the  products  measured.  By  the  usual  design  procedure,  the  response  of  the  modulator 
can  be  made  uniform  over  a  wide  range  of  frequencies. 

With  values  for  noise,  band  width,  and  modulation  which  are  readily  attained,  analyzers 
have  been  constructed  capable  of  measuring  second-order  products  80  db  down  and  third- 
order  products  100  db  down  on  their  fundamentals.  Here  a  three-stage  amplifier  was 
used  with  a  gain  of  130  db.  The  net  analyzer  gain  including  losses  in  the  modulator  and 
band  filter  was  120  db. 

Where  products  larger  in  relation  to  their  accompanying  fundamentals  are  to  be  meas- 
ured, the  amplifier  gain  may  be  reduced.  Where  products  smaller  in  relation  to  their 
fundamentals  are  to  be  measured,  it  becomes  necessary  to  insert  added  discrimination 


METHODS  OF  WAVE  ANALYSIS 


11-63 


against  the  fundamentals  in  the  input  to  the  modulator,  so  that  we  arrive  at  a  combination 
of  resonance  and  heterodyne  types. 

The  osciUator  used  for  heterodyning  is  designed  to  have  a  high  degree  of  frequency  and 
amplitude  stability  with  supply  voltage  and  temperature  variations,  and  with  'time. 
Harmonics  in  the  output  are  generally  kept  more  than  40  db  down  on  the  fundamental. 
The  oscillator  amplitude  applied  to  the  modulator  is  made  much  greater  than  the  signal 
input  and  slightly  smaller  than  the  grid  bias,  so  that  grid  current  does  not  flow  in  the 
modulator  tubes.  This  amplitude,  moreover,  should  be  maintained  fairly  constant  over 
the  frequency  range.  All  these  requirements  can  be  met  readily  with  a  bridge-type  oscil- 
lator using  thermistor  or  varistor  stabilization.  Frequency  variation  is  most  easily  effected 
when  the  oscillator  itself  is  of  the  heterodyne  type,  but  frequency  stability  is  not  as  high 
and  spurious  products  have  to  be  guarded  against. 

The  ability  of  the  analyzer  to  discriminate  against  neighboring  input  components  is 
determined  largely  by  the  properties  of  the  modulator  tubes  and  by  the  selectivity  of  the 
selective  network  of  Fig.  10,  which  may  be  made  up  of 
electromechanical,  electrical,  or  piezoelectric  elements. 
Which  of  these  is  used  is  a  matter  of  convenience  and  of 
the  frequency  at  which  the  band  is  to  be  located.  Quartz 
crystals  are  generally  used  from  40  ke  up  to  several  mega- 
cycles. The  band  width  is  made  narrow  to  cut  down  the 
noise  introduced  by  the  modulator  to  the  greatest  possible 
extent,  as  well  as  to  reduce  unwanted  components.  On 
the  other  hand,  the  band  width  should  not  be  made  so 
small  that  the  tuning-in  of  the  desired  products  becomes 
unnecessarily  difficult  and  time-consuming.  An  equiva- 
lent band  width  of  5  to  20  cycles  is  found  to  be  a  useful 
compromise  between  the  two  requirements,  the  greater 
band  width  being  used  for  measurements  at  radio  broad- 
cast frequencies.  Analysis  of  waves  in  the  short-wave 
region  requires  wider  bands  to  accommodate  incidental 
frequency  variations;  2000-cycle  bands  have  been  used. 
The  frequency  at  which  the  band  is  located  depends  upon 
the  location  of  the  frequency  spectrum  to  be  analyzed. 

In  order  to  avoid  confusion  of  the  measured  compo- 
nents with  extraneous  products  generated  in  the  modu- 
lator, the  band  is  placed  either  above  or  below  the  spec- 
trum to  be  analyzed.  Thus,  in  analyzers  designed  for  use 
at  carrier  or  radio  frequencies,  the  band  may  be  set  at  a 
low  frequency;  frequencies  ranging  from  a  fraction  of  a 
cycle  to  1000  cycles  have  been  used  in  various  types.  To 
avoid  ambiguity  in  the  measurement  of  small  power- 
supply  ripples  or  modulation  products,  the  band  should 
be  made  narrow  and  offset  from  harmonics  of  the  power 
frequency.  In  an  analyzer  covering  the  range  from  50  kc 

to  6  Me,  the  crystal  filter  is  located  at  6.7  Me  and  has  a  n  nemueiiuy,  iw  ^uca.  jj, 
band  width  of  400  cycles.  In  audio  analyzers  the  beat  piezoelectric  network,  midband  50 
,.  -  n  i  j  i_  j.t  u  j  1 1  i  -A  kc.  C,  resonant  circuit  (Q  =  100) 

frequency  is  generally  placed  above  the  band;  11  kc,  ou  tuned  to  1  kc. 

kc,  and  90  kc  have  been  used.     Figure  11  shows  the  fre- 
quency discrimination  obtainable  in  several  different  types  of  selective  circuits.     The 
100-cycle  network  referred  to  in  that  figure  is  a  cumbersome  multisection  affair  weighing 
close  to  100  Ib. 

Since  the  amplifier  is  required  to  function  over  only  the  narrow  band  passed  by  the 
selective  filter,  it  is  conveniently  built  with  screen-grid  tubes  or  pentodes  and  tuned  inter- 
stage coupling  which  permit  of  realizing  comparatively  high  gains  per  stage.  The  tuned 
interstage  coupling  supplements  the  selectivity  of  the  band  filter  and  limits  the  noise 
output  of  the  amplifier.  An  input  transformer  with  a  high  step-up  ratio  may  be  used  to 
minimize  the  effect  of  tube  noise  arising  in  the  first  amplifier  stage. 

Machine  Noise  Analyzer  Unit;  Mechanical  Band-pass  Filter.  This  device  is  used  over 
the  range  from  about  30  to  5000  cycles.  It  employs  a  mechanical  band-pass  filter  working 
at  6000  cycles  and  having  an  equivalent  band  width  of  about  25  cycles.  The  effect  of  a 
double-balanced  modulator  is  obtained  in  the  electromagnetic  structure  through  which 
the  driving  force  is  applied  to  the  filter.  In  covering  the  above  frequency  range,  the 
frequency  of  the  heterodyning  oscillator  is  varied  from  6000  to  11,000  cycles.  A  discrimi- 
nation of  about  48  db  60  cycles  from  the  tuned  frequency  is  obtained  in  the  four-section 
mechanical  filter.  The  analyzer  unit  does  not  contain  input  or  measuring  circuits  since 


band 


10     20     30     40      50     50 
Cycles  from  NCdband 

11.  Frequency  Discrimina- 
A,  electrical  network,  mid- 
frequency,  100  cycles.  B, 


11-64 


ELECTRICAL  MEASUREMENTS 


it  is  intended  to  be  used  in  conjunction  with  a  noise  meter  which  furnishes  these  circuits. 

It  is  readily  portable.  Fig- 
ure 12  gives  an  example  of 
the  noise  spectrum  of  a 
small  synchronous  motor 
obtained  with  this  ana- 
lyzer. 

General-purpose  Ana- 
lyzer Unit  with  Piezoelec- 
tric Band-pass  Filters. 
With  this  device,  it  is  pos- 
sible to  select  either  a  27- 
cycle  band  or  200-cycle 
band,  from  any  part  of  the 
frequency  range  between 
30  and  1 1 ,000  cycles.  Lat- 
tice-type quartz  crystal  fil- 
ters working  at  50  kc  are 
employed  for  both  hands. 
For  the  narrow  band,  a  dis- 
crimination of  52  db  60  cy- 
cles from  the  center  of  the 


DU 
55 
^50 

CO 

§45 

i«> 

^35 

Iso 

,0 

•S3 
25 

20 

)      200    400    600    800    100O  1200  1400  1600  180020002200  24C 
frequency  irt  Cycles  per  Second 

FIG.  12. 


Noise  Analysis  of  Small  Synchronous  Motor — 1/4  hp,  1800 
rpm,  60  cycle,  three  phase,  220  volts,  no  load 


band  is  obtained,  while  the  wide  band  has  a  similar  value  350  cycles  from  midband.  The 
circuit  includes  a  demodulator  by  means  of  which  the  wave  components  selected  by  the  fil- 
ter are  translated  to  their  original  frequencies 
as  indicated  in  the  schematic  of  Fig.  13.  This 
^*  arrangement  is  particularly  valuable  in  mea- 
suring closely  spaced  products,  since  it  avoids 
output  indication  produced  by  transmission 

of  the  heterodyne  source  itself  through  modu- 

FIG.  13.  Heterodyne  Analyzer  Circuit  Used  for  lator  unbalance  (carrier  leak).  The  modula- 
Frequency  Selection.  The  modulator-demodu-  tor  leak  here  is  translated  to  zero  frequency 
JMSS&7S:  Sffi383£Z££  ^d  so  is  not  transmitted  by  the  a-c  system 

connected  to  the  demodulator  output.    The 

demodulator  leak  is  far  removed  in  frequency  from  the  selected  component  and  can  be 
made  harmless.  Both  modulator  and  demodulator  are  of  the  double-balanced  copper  oxide 


J.UU 

90 
080 

or 
05 

<5 

570 

I 

"t  60 

O3 

§ 

JD 

o50 

**S 

40 

30 

( 

J^ 

•^ 

1 

-L-^__ 

-U^L. 

V 

fifh  no  So 
Fan  a 

ind  Insulat 
id  Exhaust 

on  on 

\_ 

"  L. 

\ 

r-r 

-r~U 

I 

lr 

Fan  and  E 
sulated  fo 

haust  —  * 
Noise 

V 

I-, 

-u-^ 

_J—  — 

^ 

-L^ 

*-I-^_ 

"-u^ 

>          1000       2000       3000       4000       5000       6000       7000       8000       9000     10,000 
Frequency  -  Cycles  per  Second 
FIGL  14.     Band-frequency  Analysis  of  Exhaust  Fan  Noise 

SPECTROGRAPHS 


11-65 


type  and  are  supplied  with  carrier  heterodyning  power  from  the  same  oscillator  through 
buffer^  amplifier  stages.  The  analyzer  unit  does  not  include  input  or  measuring  circuits 
S1??VVS  mtended  as  an  attachment  for  a  noise  meter  or  similar  measuring  device  of 
which  these  circuits  are  a  necessary  part.  Figure  14  gives  an  example  of  the  use  of  the 
200-cycle  band  in  the  analysis  of  exhaust  fan  noise  to  determine  the  noise  reduction  ob- 
tained at  various  frequencies  by  means  of  sound  insulation  on  the  fan  and  exhaust. 


31.  SPECTROGRAPHS 

Investigation  of  the  spectrum  of  a  complex  wave  by  the  methods  just  presented  is  a 
comparatively  slow  process  at  best  since  manual  adjustment  of  a  resonance  frequency  or 
of  a  beating  oscillator  frequency  is  required  to  shift  the  analyzer  from  point  to  point  along 
the  band  to  be  studied.  To  speed  up  the  process  the  analyzer  may  be  tuned  automatically, 
and  the  spectral  distribution  displayed  in  its  entirety. 

RECORDING  TUNED  CIRCUIT  ANALYZER.  Here  the  resonant  frequency  of  a  single 
electrical  tuned  circuit  is  varied  in  small  steps  over  the  frequencj-  range  by  means  of  a 
player-piano  pneumatic  control.  Simultaneously  a  photographic  record  is  made  of  the 
current  in  the  resonant  circuit  after  amplification  and  rectification,  the  photographic  paper 
being  correspondingly  stepped  in  position  to  accord  with  the  frequency  scale.  Two  fre- 
quency ranges  are  provided:  20-1250  cycles  and  SO-5000  cycles.  The  complete  recording 
takes  about  5  minutes.  A  20-db  amplitude  range  is  the  useful  limit  for  a  single  record, 
the  amplitude  scale  on  the  photographic  record  being  approximately  linear.  Discrimina- 
tion in  the  80-5000  cycle  band  is  about  20  db  60  cycles  away  from  the  tuning  frequency, 
decreasing  somewhat  when  the  tuning  frequency  is  above  1000  cycles.  Figure  15  presents 


100 


400  600  100O»  2000 

Fro.  15.     Record  of  160-cycle  Buzzer  Output 


300O  5000 


the  analysis  of  a  buzzer  tone,  each  peak  of  the  curve  denoting  the  amplitude  and  frequency 
of  an  input  component. 

In  any  scheme  of  this  kind  which  shifts  a  single  discriminating  circuit  over  the  bandT  a 
definite  compromise  has  to  be  made  between  the  speed  at  which  the  circuit  is  shifted  and 
the  consequent  resolving  power,  or  ability  to  separate  neighboring  components.  This 
relationship  is  discussed  quantitatively  below  in  the  discussion  of  the  Sweep  Frequency 
Heterodyne. 

TUNED-REED  ANALYZER.  This  analyzer  makes  use  of  a  series  of  eleetromagnetically 
driven  reeds  tuned  to  frequencies  in  the  audio  range,  distributed  with  a  uniform  percentage 
difference  in  frequency.  Each  reed  carries  a  small  concave  mirror  which  deflects  a  beam 
of  light  by  an  amount  proportional  to  the  oscillation  amplitude  of  the  reed.  The  vibrating 


Cycles  per  Second 
400 


800 


1600 


3200 


Piano  Tone 


Piano  Tone  F 
FIG.  16.     Frequency  Spectra  Obtained  with  Reed  Analyzer 

spots  of  light  indicate  on  a  ground-glass  screen  the  components  present  in  the  input  wave 
which  is  applied  to  the  electromagnets  driving  the  reeds.  Damping  of  the  reeds  is  pro- 
portional to  frequency  and  provides  a  discrimination  of  20  db  for  components  2  1/2  per  cent 
from  the  tuned  frequency.  A  component  20  db  down  on  the  maximum  can  be  evaluated 


11-66 


ELECTRICAL  MEASUREMENTS 


with  an  accuracy  of  about  1  db  when  its  frequency  coincides  with  that  of  one  of  the  reeds. 
A  permanent  record  of  a  spectrum  distribution  constant  in  time  can  be  obtained  photo- 
graphically. To  record  wave  components  varying  with  time  a  motion-picture  camera  must 
be  used.  Figure  16  shows  spectrograms  for  two  piano  tones  obtained  with  a  demonstration 
analyzer  of  this  type,  covering  the  frequency  range  from  50  to  3200  cycles  with  144  reeds. 
COMMUTATED  BAND  ANALYZER.  Selection  of  components  is  effected  by  a  bank 
of  contiguous  fixed  band  niters  extending  over  the  frequency  range  to  be  covered.  The 
output  of  each  band  filter  is  rectified,  filtered,  and  amplified,  as  shown  in  Fig.  17.  The  ver- 
tically deflecting  (V)  plates  of  a  cathode-ray  tube  are  connected  to  each  channel  m  turn 
by  means  of  a  commutator,  while  the  horizontally  deflecting  (H)  plates  are  connected  to  a 
synchronously  operated  stepped  sweep.  In  this  way,  a  linear  frequency  scale  is  laid  out 
along  a  horizontal  axis  on  the  cathode-ray  screen,  and  a  vertical  line  represents  the  wave 
amplitude  or  energy  within  the  corresponding  band  filter.  The  general  aspect  of  spectro- 
grams of  this  type  is  indicated  on  the  oscilloscope  screen  depicted  in  the  figure. 


FIG.  17.     Commutated  Band  Analyzer;  Cathode-ray  Presentation 

From  10  to  30  filters  have  been  used  with  band  widths  distributed  linearly  or  logarithmi- 
cally to  cover  the  audio  spectrum.  With  the  linear  distribution,  bandwidths  of  300  to  150 
cycles  have  been  used.  Filter  attenuations  at  the  crossovers  are  3  to  6  db,  and  at  the  mid- 
band  frequencies  of  the  immediately  adjacent  niters  are  of  the  order  of  20  db.  A  non- 
linear compressor  inserted  in  the  lead  to  the  vertically  deflecting  (T7)  plates  of  the  cathode- 
ray  tube  helps  to  make  the  smaller  amplitudes  more  readily  perceptible.  Accurate  por- 
trayal of  an  extended  amplitude  range,  however,  requires  increase  in  filter  selectivity  to 
reduce  interference  from  components  falling  within  the  attenuating  regions  of  the  filters. 
Commutation  should  be  fast  enough  to  reveal  any  envelope  variations  transmitted  by  the 
band  filters.  Thus  if  the  widest  filter  band  of  the  bank  is  B  cycles  wide,  variations  at  a 
rate  of  B/2  are  freely  transmitted.  The  rectifier  low-pass  filter  therefore  should  be  B/2 
cycles  wide,  or  a  little  wider.  For  faithful  indication,  then,  each  channel  should  be  sampled 
B  times  per  second  at  least.  While  mechanical  commutators  have  been  shown  for  sim- 
plicity in  Fig.  17,  the  channel  sampling  commutator  may  be  replaced  by  electronic  means, 
including  clamps  and  a  ring,  with  as  many  stages  as  there  are  bands  to  sample.  Similarly 
the  sweep  commutator  may  be  replaced  by  a  properly  synchronized  electronic  sweep 
developing  a  sawtoothed  wave  form. 

SWEEP  FREQUENCY  HETERODYNE,  CATHODE-RAY  PRESENTATION.  This 
method  uses  what  is  essentially  an  automatic  heterodyne  analyzer  and  is  in  general  use 
throughout  the  radio  frequency  spectrum,  from  the  broadcast  band  and  below,  to  the 
centimeter  region.  A  block  diagram  of  the  circuit  arrangement  is  shown  in  Fig.  IS.  There 
a  sweep  generator  provides  a  sawtooth  wave  at  a  sub-audio  repetition  rate  for  two  purposes. 
First,  it  constitutes  the  horizontal  sweep,  and  second,  it  provides  a  means  for  varying  the 
beating  oscillator  frequency  throughout  a  definite  frequency  range.  To  accomplish  the 
second  function,  the  sawtooth  wave  actuates  a  reactance  tube  associated  with  the  fre- 
quency-determining circuit  of  the  oscillator.  Or,  at  ultra-high  frequencies,  the  sawtooth 
wave  is  impressed  directly  upon  an  oscillator  tube  of  the  velocity- variation  type.  In 
either  case,  the  oscillator  frequency  is  made  linearly  proportional  to  the  instantaneous 
amplitude  of  the  sweep  throughout  its  utilized  portion. 

The  variable  oscillator  output  then  modulates  the  input  wave  to  be  analyzed  so  as  to 
sweep  its  spectrum  across  the  narrow  pass  band  of  the  i-f  amplifier.  The  i-f  output  is 
tiien  detected,  amplified,  and  applied  to  the  V  plates  of  a  cathode-ray  oscilloscope. 


SPECTBOGKAPHS 


11-67 


Sweep-frequency  Heterodyne  Analyzer;  Cathode-ray  Pres- 
entation 


The  amount  of  frequency  variation  (F)  during  a  sweep  cycle  is  made  large  enough  to 
include  the  region  to  be  analyzed.  The  sweep  period  (T)  and  the  i-f  band  width  (B)  must 
be  selected  to  accord  with  the  resolving  power  required.  If,  for  example,  components  equal 
in  magnitude  and  separated  by  S  cycles  are  to  be  displayed  as  distinct  pips  on  the  oscillo- 
scope screen,  the  response  produced  by  one  of  them  must  have  built  up  and  decayed  to  a 
sufficiently  low  value  before  the  next  component  enters  the  i-f  band.  The  transient 
response  depends  upon  the  i-f  band  width  and  has  a  duration  of  roughly  %/B  seconds.  This 
time  is  to  be  no  greater 
than  the  time  required  for 
the  sweep  to  traverse  the 
frequency  spacing  between 
the  two  components,  or 
ST/F.  If  then  the  i-f 
bandwidth  is  made,  say, 
one-quarter  the  frequency 
separation  between  the 
components  to  be  resolved, 
the  sweep  period  T  should 
be  of  the  order  of  magni- 
tude F/2£2.  Where  the  FlG-  18* 
components  to  be  resolved 
are  not  equal  in  magnitude,  the  sweep  period  or  the  sweep  frequency  deviation  must  ^  be 
correspondingly  reduced,  if  the  two  transients  are  not  to  overlap  excessively.  Resolution 
of  a  carrier  and  two  sidebands  is  indicated  on  the  oscilloscope  screen  of  Fig.  18.^ 

In  certain  cases  involving  large  numbers  of  components,  it  is  desirable  to  display  the 
envelope  of  the  spectral  distribution  rather  than  to  resolve  the  components  individually. 
There  the  i-f  band  (B}  is  made  wide  enough  to  include  a  number  of  components,  and  the 
approximate  relations  given  above  remain  applicable  when  the  separation  (S)  refers  to 
bands  rather  than  to  individual  components.  An  example  is  shown  in  Fig.  19  of  a  spectrum 
envelope  applying  to  a  magnetron  pulsed  at  1SOO  cps,  taken  with  an  i-f  band  width  of  the 
order  of  50  kc.  • 

Various  applications  involve  ranges  of  T  from  1/100  to  perhaps  1/2  second,  of  #  from 
2  to  100  kc,  and  of  F  from  100  kc  to  200  Me. 

One  of  the  most  satisfactory  methods  for  calibrating  the  frequency  scale  is  to  superpose 
on  the  modulator  input  of  Fig.  IS  a  standard  frequency  source  which  is  amplitude  modu- 
lated by  a  sine  wave  at  a  known  and  adjustable  rate. 
This  superposes  three  pips  at  known  frequencies  ^on 
the  signal  spectrum,  which  can  be  varied  in  position 
to  coincide  with  points  of  interest. 

For  presentation  of  individual  components  a  linear 
rectifier  may  be  used  to  provide  a  linear  output  scale. 
For  envelopes  a  square-law  rectifier  provides  an  out- 
put proportional  to  energy.  Measurement  of  output 
levels  may  be  carried  out  by  one  of  two  methods. 
The  deflections  may  be  read  on  a  calibrated  scale  on 
the  cathode-ray  screen,  or  a  calibrated  attenuator  in 
either  input  or  i-f  paths  may  be  varied  to  bring  to  a 
fixed  deflection  the  particular  part  of  the  distribution 
which  is  to  be  evaluated.  In  either  method,  over- 
loading of  the  measuring  system  must  be  avoided. 
A  similar  sweep  type  of  heterodyne  analyzer,  de- 
veloped primarily  for  the  study  of  musical  tones,  has 
a  galvanometer  as  output  indicator  A  rotating 
mirror  synchronized  with  the  10-cycle  oscillator  fre- 
quency sweep  reflects  the  galvanometer  light  spot  to 
a  ground-glass  screen.  The  rotating  mirror  thus  pro- 
vides the  equivalent  of  a  horizontal  sweep.  An 
electrical  network  is  used  at  20  kc  with  a  band  width  sufficient  to  resolve  tones  separated 

YAnother  indicating  and  recording  device  used  in  conjunction  with  the  sweep  frequency 
heterodyne  was  developed  primarily  for  noise  and  vibration  studies  in  the  audio  band  It 
uses  a  scriber  provided  with  a  synchronous  mechanical  drive  (like  that  of  Fig  20)  which 
furnishes  a  permanent  record  of  the  spectral  distribution  upon  waxed  paper  Piezo  niters 
make  available  analyzer  band  widths  of  5,  50,  and  200  cycles.  The  amplitude  scale  covers 
a  range  of  80  db.  A  complete  record,  directly  legible,  requires  about  2  minutes  time. 


7  me 

FIG.  19.     Spectrogram  of  an  Oscillating 

Magnetron    Pulsed    at    1800—;    Pulse 

Duration    Roughly    0.25    Microsecond. 

Wave  components  are  not  resolved. 


11-68 


ELECTRICAL  MEASUREMENTS 


SOUND  SPECTROGRAPH.  The  sound  spectrograph  produces  a  visual  record  show- 
ing the  distribution  of  energy  within  an  audio  band  in  both  frequency  and  time.  Though 
the  development  of  this  device  must  be  regarded  as  still  in  the  experimental  stage,  it  con- 
stitutes as  it  stands  a  powerful  means  for  the  analysis  of  speech,  music,  and  noise.  Its 
power  comes  from  the  high  concentration  of  information  presented,  which  permits  details 
of  the  spectral  distribution  to  be  followed  as  a  function  of  time.  _ 

Figure  20  shows  a  simplified  schematic  of  one  form  of  the  device  which  will  serve  to 
illustrate  the  basic  idea.  Three  distinct  functions  are  involved.  First  the  sound  to  be 
analyzed  is  recorded  so  that  it  can  be  repeatedly  reproduced.  Here  a  magnetic  tape 
recording  is  shown,  mounted  on  a  rotating  disk  which  is  driven  by  a  synchronous  motor. 
Second,  analysis  of  the  recorded  sound  is  effected  by  a  heterodyne  type  of  analyzer  in 
which  the  oscillator  frequency  is  varied  to  move  the  analyzer  filter  in  effect  steadily  over 
the  sound  spectrum.  This  variation  is  indicated  in  the  figure  by  mechanical  coupling 
between  the  varying  condenser  of  the  beating  oscillator  and  the  main  drive  shaft.  In 
this  way  the  analyzing  frequency  changes  a  small  amount  throughout  each  revolution  of 
the  shaft.  Finally  the  analyzer  output  is  recorded  in  synchronism  with  the  ^reproduced 
sound.  Recording  is  accomplished  in  the  same  manner  as  that  practiced  in  facsimile 

reception.  A  drum  bearing 
the  recording  paper  is  coupled 
to  the  main  drive  shaft,  and 
the  recording  stylus  is  moved 
laterally  a  small  distance  in 
each  revolution  by  means  of  a 
lead  screw  similarly  coupled. 
The  electrically  sensitive  paper  ' 
mounted  on  the  drum  is 
marked  by  the  stylus  with 
gradations  of  density  which 
accord  with  the  analyzer  out- 
put. The  net  result  is  to  pro- 


duce    a    sort    of 
sional    picture 


three-dimen- 
which   the 


FIG.  20.  Schematic  Representation  of  One  Form  of  Sound  Spec-  energy  distribution  is  depicted 
trograph.  The  output  of  the  sweep-frequency  heterodyne  ana-  v  d  it  variations  on  a 
lyzer  is  displayed  as  a  pattern  on  electrically  sensitive  paper,  ex-  "J  •*  .  , 

hibiting  the  energy-frequency  distribution  as  a  function  of  time,    rectangular  plot   of  frequency 

against  time. 

A  signal  level  range  of  35  db  is  handled  with  the  aid  of  automatic  volume  control  which 
provides  something  like  a  threefold  compression  on  a  db  basis.  Two  heterodyne  band 
widths  are  available:  one  45  and  the  other  300  cycles,  as  measured  at  the  3-db  points.  The 
linear  frequency  scale  of  the  spectrogram  is  2  in.  high  and  covers  the  range  from  100  bo 
3500  cycles.  The  linear  time  scale  extends  for  a  length  of  12  in.,  corresponding  to  an 
original  audio  sample  2.4  seconds  in  duration.  Two  hundred  rotations  of  the  disk  carrying 
the  audio  sample  are  completed  in  less  than  5  minutes  for  the  full  analysis.  The  most 
recent  development  in  this  field  speeds  up  the  analysis  by  a  factor  of  the  order  of  200. 

Photographic  reproductions  of  spectrograms  of  speech,  music,  and  noise  are  to  be  found 
in  the  last  four  references  listed  under  Spectrographs  in  the  Bibliography.  In  the  case  of 
speech,  the  narrow  band  (45  cycles)  analysis  is  adequate  to  resolve  individual  harmonics 
of  the  voiced  sounds.  The  traces  curve  up  and  down  as  the  pitch  of  the  voice  varies,  and 
the  spacing  between  harmonics  gets  bigger  as  the  pitch  increases  at  any  particular  instant. 
A  definite  loss  of  detail  occurs  when  the  wide  baud  (300  cycles)  is  used  since  two  or  three 
harmonies  are  merged.  A  spectrogram  of  thermal  noise  shows  energy  concentrated  in 
different  frequency  regions  at  different  instants  of  time.  Randomly  spaced  vertical  spin- 
dles of  the  spectrogram  correspond  in  length  to  the  300-cycle  filter  of  the  analyzer.  Espe- 
cially interesting  are  spectrograms  of  a  warble  tone  which  constitutes  a  frequency-modu- 
lated wave,  produced  by  varying  sinusoidally  the  frequency  of  an  oscillator.  The  narrow 
filter  reveals  the  presence  of  individual  side  frequencies  of  a  tone  warbled  at  a  50-cycle 
rate,  but  the  wide  filter  cannot  resolve  them;  they  are  integrated  to  reveal  the  instantaneous 
frequency. 

The  process  of  sound  portrayal  employed  in  the  sound  spectrograph  described  above 
results  in  a  considerable  simplification  of  apparatus  where  high  resolution  is  desired.  The 
equivalent  machine  to  record  high-resolution  patterns  directly  would  require  something 
of  the  order  of  a  hundred  filters  rather  than  a  single  one.  Sound  spectrographs  of  this  type 
have  been  made  to  transcribe  long  samples  of  sound  by  employing  the  disk  and  drum 
arrangement  of  Fig.  20  with  endless  belts,  one  of  magnetic  tape  and  another  of  recording 
paper. 


MICROWAVE   MEASUREMENTS  11-69 

For  special  uses,  such  as  experimental  visual  telephony  for  the  deaf,  low  resolution  pat- 
terns of  the  type  provided  by  the  sound  Spectrograph  are  formed  at  speech  rates.  Ten  to 
twenty  fixed  filters  or  a  scanning  band  affording  equivalent  resolution  are  used  in  this 
instrument,  termed  "visible  speech  translator."  The  outputs  produce  speech  patterns  in 
light  upon  a  moving  band  of  either  phosphorescent  or  fluorescent  material.  Scanning 
and  timing  functions  are  controlled  by  synchronized  electronic  means.  The  accuracy  and 
speed  of  these  translators  are  sufficient  to  permit  their  use  for  the  reading  of  continuous 
speech  by  trained  observers. 

BIBLIOGRAPHY 

Resonance  Analyzers 

^^^  ^^BAyrf^n?lectrical  ^ave  Analyzers  for  Power  and  Telephone  Systems,  Trans.  AJ.B.B., 
Vol.  48,  Il\j7  (1929). 

Sivian,  Dunn,  and  White,  Absolute  Amplitudes  and  Spectra  of  Certain  Musical  Instruments  and 

Orchestras,  J.  Acous.  Soc.  Am.,  Vol.  2,  330  (1931). 
Scott,  Analyzer  for  Sub-audible  Frequencies,  /.  Acous.  Soc.  Am.,  VoL  13,  360  (1942). 

Suppression  and  Intennodulatfon  Analyzers 

Wolff,  A-c  Bridge  as  a  Harmonic  Analyzer,  J.  Optical  Soc.  Am.t  VoL  15,  163  (1927). 
•q^p  FV  Sc°2    e'  Analysis  and  Measurement  of  Distortion  in  Variable  Density  Recording,  J. 

Billiard,  Distortion  Tests  by  the  Intel-modulation  Method,  Proc.  I.R.B.,  VoL  29,  614  (1941). 
Hayes,  A  New  Type  of  Practical  Distortion  Meter,  Proc.  I.R.E.,  Vol.  31,  112  (1943). 
Pickering,  Measuring  Audio  Intel-modulation,  Electronic  Industries,  June  1946. 

Warren  and  Hewlett,  Analysis  of  the  Intel-modulation  Method  of  Distortion  Analysis,  Proc.  I.R.E., 
Vol.  36,  No.  4,  p.  457. 

Heterodyne  Analyzers 


Vol.  18,  No.  1,  178  (1930). 
Arguimbau,  Wave  Analysis,  Gen.  Radio  Experimenter,  VoL  8,  Nos.  1  and  2,  12  (1932), 
Castner,  A  General  Purpose  Frequency  Analyzer,  Bdl  Labs.  Rec.,  VoL  13,  267  (1935). 

Spectrographs 

Wegel  and  Moore,  An  Electrical  Frequency  Analyzer,  Bell  Sys.  Tech.  J.,  VoL  3,  299  (1924). 

Potter,  Transmission  Characteristics  of  a  Short  Wave  Telephone  Circuit,  Proc.  I.R.E.,  VoL  18,  583 

(1930). 

Hickman,  An  Acoustic  Spectrometer,  J".  Acous.  Soc.  Am.,  VoL  6,  108  (1934). 
Wolf  and  Sette,  Some  Applications  of  Modern  Acoustic  Apparatus,  J.  Acous.  Soc.  Am.,  VoL  6,  160 

(1935). 
Schuck,  The  Sound  Prism,  Proc.  I.R.E.,  VoL  22,  1293  (1939) ;  Panoramic  Reception,  Electronics,  VoL 

14,  36  (1941);  Recording  Sound  Analyzer,  Electronics,  VoL  16,  100  (July  1943). 
Williams,  R.F.  Spectrum  Analyzers,  Proc.  I.R.E.,  January  1946. 

Gaffney,  A  Spectrum  Analyzer  for  Microwave  Pulsed  Oscillators,  Waves  and  Electrons^  Vol.  1,  83  (1946). 
Apker,  Kahnke,  Taft,  and  Watters,  Wide  Range  Double  Heterodyne  Spectrum  Analyzers,  Proc.  I.R.B., 

1947. 

Montgomery,  Techniques  of  Microwave  Measurements,  Rad.  Lab.  Series  VoL  11,  Chapter  7,  McGraw- 
Hill. 
Koenig,  Dunn,  and  Lacy,  The  Sound  Spectrograph;  Riesz  and  Schott,  Cathode  Ray  Translator; 

Dudley  and  Gruenz,  Visible  Speech  Translators  with  External  Phosphors,  J.  Acous.  Soc.  Am., 

July  1946. 

Potter,  Kopp,  and  Green,  Visible  Speech,  Van  Nostrand,  1946. 
Kersta,  Amplitude  Cross-section  Representation  with  the  Sound  Spectrograph,  J.  Acous.  Soc.  Am., 

November  1948. 
Mathes,  Norwine,  and  Davis,  The  Cathode  Ray  Sound  Spectroscope.    J.  Acous.  Soc.  Am.,  September 

1949. 

General  Surveys 

Bourne,  Wave  Analysis,  Electronic  Eng.t  VoL  15,  149,  281,  472;  VoL  16,  31  (1942);  3,  bibliography. 
Scott,  Measurement  of  Audio  Distortion,  Communications,  VoL  26,  No.  4,  23  (1946). 


MICROWAVE  MEASUREMENTS 

By  E.  W.  Houghton 

The  electrical  characteristics  of  any  network  can  be  completely  expressed  in  terms  of 
three  quantities:  impedance,  power,  and  frequency.  Attenuation  ratio,  although  not  a 
fundamental  quantity,  is  of  sufficient  importance  to  warrant  separate  consideration.  This 
section  is  concerned  with  methods  and  techniques  for  accurately  measuring  these  quan- 
tities in  the  frequency  range  of  roughly  1000-30,000  megacycles.  The  methods  can  be 
used  to  attain  the  accuracies  summarized  in  Table  1. 


11-70 


ELECTRICAL  MEASUREMENTS 


Table  1.     Microwave  Measurement  Accuracies 


Quantity  to  Be  Measured 

Method 

Accuracy 

Impedance: 

Slotted  line 

±2% 

Directional  coupler 
Hybrid  junction 
Slotted  line 

±5%  -  ±0.2% 
±5%  -  ±0.2% 
±5% 

Power  (averaged  over  several  seconds) 
5-200  microwatts     

Bolometer 

±0.5db 

02  10  milliwatts                

Bolometer 

±0.1  db 

00  1-1  00  watts          

Bolometer-attenuator 

±0.3db 

20-  1  00  watts  

Calorimeter 

±0.3db 

Power  (averaged  over  several  microseconds) 
1-  1  00  watts           

Calibrated  crystal 

±0.6  db 

0  1—100  kilowatts                       

Calibrated  crystal 

±1  db 

Attenuation 
0-3  db      

Bolometer 

±0.05db 

0-  1  3  db    

Bolometer 

±0.1  db 

0_60  db                                    

Heterodyne  receiver 

±0.2db 

0-40  db           .              

R-f  attenuator 

±0.2db 

Frequency: 
Small  differentials          .        .-• 

Wavemeter 

,  ±0.005% 

Absolute          

Harmonic  generator 

±0,0001% 

Wavemeter 

±0.01% 

32.  IMPEDANCE  MEASUREMENTS 

TRANSMISSION-LINE  CALCULATIONS.  At  microwave  frequencies  practically  all 
measurable  impedances  are  in  transmission  lines  (coaxial  or  wave  guide)  ,  and  the  terminals 
of  these  impedances  are  transmission-line  couplings  (i.e.,  coaxial  jacks  and  plugs  or  wave- 
guide cboke  or  plane  flanges)  or  simply  specific  transverse  planes  in  the  transmission  line 
(see  reference  17,  p.  11-89).  Absolute  impedance  values  are  rarely  important,  and  in 
fact  wave-guide  transmission-line  impedances  can  be  denned  in  several  ways  (see  refer- 
ence 1).  Ambiguity  is  removed  by  using  a  relative  or  "normalized"  value  which  is  the 
ratio  of  an  impedance,  2,  to  ZQ,  the  characteristic  impedance  of  the  transmission  line  defined 
in  the  same  way;  thus  z  =  Z/Z^  where  z  is  a  complex  value. 

A  section  of  uniform,  lossless,  transmission  line,  Fig.  1,  transmits  energy  in  a  given 
mode  between  a  source  plane,  s-s,  and  an  impedance  plane,  t-t,  by  means  of  traveling 
electromagnetic  waves.  Unless  the  impedance  at  t-t  termi- 
nates the  line  in  its  characteristic  impedance,  two  such  waves 
exist:  (1)  an  incident  wave  which  originates  at  the  source  and 
travels  forward  toward  the  impedance  plane  t-t,  and  (2)  a 
reflected  wave  which  originates  at  t-t  and  travels  backward 
toward  the  source  (see  reference  2).  The  ratio  of  transverse 
voltage  in  the  reflected  wave  to  transverse  voltage  in  the 
incident  wave  defines  the  reflection  coefficient  k,  which  is  a 
complex  quantity.  Total  transverse  voltage  and  longitudinal 
current  on  the  transmission  line  are  the  vector  sums  of  the 
voltages  and  currents  in  the  two  oppositely  traveling  waves,  which  therefore  combine  to 
produce  a  standing-wave  pattern  distributed  along  the  transmission  line.  The  ratio  of 
the  maximum  to  the  minimum  total  voltage  defines  (voltage)  standing-wave  ratio,  S, 
which  is  not  a  complex  quantity. 

The  more  general  result  of  interference  between  oppositely  traveling  waves  is  to  trans- 
form the  impedance  at  t-t  to  new  values  at  planes  between  s-s  and  t-L  The  relationship 
between  reflection  coefficient  and  normalized  impedance  at  t-^t  is  :  Kt  \4>  =  (zt  —  l)/(z<  +  1), 
where  zt  is  a  complex  value.  At  some  arbitrary  plane  a-a  closer  to  the  source  and  removed 
from  t-t  by  line  length  l/\g  wavelengths  the  reflection  coefficient  is  ka  =  Kt  \<j>  —  20, 
where  &  -  2?rZ/Xg  radians.  The  normalized  impedance  at  this  plane  is  thus: 

1  -f 


9    Incident 
jvave 

o       b   < 

j 

i  t 
1 

I 
1 

t 

! 

Reflected 
s      wave       < 

,   ! 

;        b  < 

1        " 
1    * 

FIG,  1.     Impedance  Planes  on 
a  Transmission.  Line 


This  car*  be  expressed  as  &*  =  (st  +  j  tan  #)/(!  -+-  jzt  tan  fl). 

The  oppositely  traveling  waves  combine  in  phase  opposition  to  give  a  minimum  voltage 


IMPEDANCE  MEASUREMENTS 


11-71 


standing  wave  at  2<r  «  (»  -  <£)  radians,  or  (TT  -  £)/47r  wavelengths  from  «;  at  */2 
radians  or  1/4  wavelength  closer  to  s-s  the  waves  directly  add  to  give  a  maximum  voltage 
standing  wave.  The  standing-wave  ratio  is  thus  8  =  (1  +  j£*)/(l  —  Kt}.  From  measure- 
ments of  either  K,  or  S,  and  o-,  the  normalized  impedance  at  t-4  can  be  computed  (see 
reference  3)  from 

1  - 


^  The  transmission-line  calculator  (see  reference  4),  Fig.  2,  provides  a  convenient  graphical 
aid  to  visualization  and  computation  of  the  above  impedance  transformations.    An  ex- 


Pi  Q.  2.    Transmission  Line  Calculator 

ample,  drawn  on  the  chart,  will  illustrate  its  use.  As  the  line  length  between  t-t  and  a-a 
increases,  transformed  impedance  values  are  intercepted  by  the  circle  in  the  order  indicated 
by  the  arrow  on  its  periphery.  Thus  a  normalized  impedance  zt  =  Q.do  -f-  j'0.45  at  t-t 
transforms  to  za  —  1.5  -f  ;0.7  at  plane  a-a  located  0.102  wavelength  from  i-i.  Normalized 
admittance  values  equivalent  to  series  impedance  values  appear  at  points  on  the  circle 
diametrically  opposite;  thus  ya  =  1/Za  =  0.55  —  J0.25.  At  b-b,  zt  is  transformed  to  a 
pure  resistance  Zb  —  2.0  +  j"0;  at  c-c,  zt  is  transformed  to  zc  =  0.5  +  ^"0,  also  a  pure  resist- 
ance. Planes  6— b  and  c~c  are  the  positions  of  the  maximum  and  minimum  standing  wave 
voltages  respectively,  and  the  standing-wave  ratio  created  by  2*  is  S  =  25  =  l/z«  =  2. 
Following  the  above  example  in  reverse  order  it  is  seen  that  the  chart  provides  an  ex- 
tremely practical  method  for  computing  the  impedance  zt  from  measurements  which-  gave 
values  of  5  -  2  and  <r  =  0.408.  By  this  method  eq.  (1)  can  be  solved  rapidly  and  with  a 
precision  adequate  for  most  measurements. 

The  results  of  impedance  measurements  are  primarily  used  to  study  the  electrical  nature 
of  the  impedance,  or  to  design  networks  for  transforming  this  impedance  to  a  new  value 


11-72  ELECTRICAL  MEASUREMENTS 

(see  reference  5)  (usually  the  characteristic  impedance  of  the  transmission  line,  which  it 
then  terminates  without  reflection).  However,  in  many  cases  a  given  impedance  may  be 
used  to  terminate  an  electrically  long  transmission  line  or  a  line  of  unspecified  length. 
Under  these  conditions  a  measurement  of  the  reflection  coefficient  phase  angle  would  be 
trivial;  thus  in  many  practical  cases  impedance  measurements  are  concerned  only  with 
ascertaining  reflection  coefficient  amplitudes  or  standing-wave  ratios  produced  by  that 
impedance  in  a  connected  transmission  line. 

STANDING- WAVE  DETECTORS.  Of  the  three  instruments  most  commonly  used 
for  impedance  measurement  the  standing-wave  detector  is  the  most  versatile  since  it 
affords  information  on  both  amplitude  and  phase  of  the  reflection  coefficient.  It  comprises : 
(1)  a  section  of  uniform  transmission  line  (coaxial  or  wave-guide)  which  has  a  longitudinal 
slot  in  its  outer  conductor,  and  (2)  a  probe  which  projects  through  this  slot  and  travels 
parallel  to  the  axis  of  the  line  on  a  carriage.  A  standing-wave  pattern  of  voltage  is  set  up 
in  the  line  when  the  test  impedance  is  connected  at  one  end  and  a  source  at  the  other. 
The  probe,  excited  by  this  voltage,  is  connected  to  a  detector  whose  output  is  a  relative 
function  of  total  voltage  amplitudes  along  the  slotted  line.  From  readings  obtained  at 
the  maximum  and  minimum  voltage  positions,  the  standing-wave  ratio  can  be  calculated, 
provided  that  the  law  of  the  detector  is  known  or  the  detector  has  been  calibrated.  The 
detector  may  be  a  crystal  (reference  6),  bolometer  (reference  8),  or  the  mixer  of  a  hetero- 
dyne receiver  (reference  25).  In  the  last  case  the  mixer  may  be  kept  linear  by  operating 
at  low  levels;  the  second  detector  is  used  at  a  constant  level  by  means  of  a  calibrated  i-f 
attenuator.  The  law  of  a  bolometer  detector  and  its  bridge  circuit  can  be  obtained  accu- 
rately by  a  low-frequency  calibration.  Specially  selected  crystals  can  be  found  to  meet  a 
square-law  requirement  over  restricted  input  ranges,  but  they  should  preferably  be  cali- 
brated against  a  bolometer  power  meter  or  heterodyne  receiver. 

A  somewhat  less  accurate  calibration  method  is  to  terminate  the  slotted  line  in  a  per- 
fectly reflecting  short  and  measure  the  standing-wave  voltage  distribution,  which  under 
ideal  conditions  approaches  V  =  Csin  (2irl/\g),  where  C  is  a  constant  and  l/\g  is  the 
displacement  in  wavelengths  from  a  plane  of  zero  voltage.  A  method  for  which  the  law 
need  not  be.  known  is  to  operate  the  detector  at  a  constant  level  by  using  a  calibrated  r-f 
attenuator  between  the  slotted  line  and  the  source,  or  between  the  pick-up  probe  and  the 
detector;  standing-wave  ratios  are  deduced  from  attenuator  settings.  The  r-f  attenuator 
method  is  satisfactory  for  measurement  of  high  standing-wave  ratios  (10-100)  but  is 
generally  considered  inferior  to  other  methods  for  low  ratios  (1  to  2) . 

In  order  to  avoid  reflection  errors  from  the  probe,  its  insertion  is  limited  to  small  values 
(less  than  10  per  cent  of  wave-guide  height  or  */4  the  difference  between  coaxial  conductor 
diameters)  for  which  its  power  output  will  be  20-30  db  lower  than  the  power  in  the  slotted 
line.  The  most  convenient  oscillator  sources  are  single-cavity  klystron  (reference  10)  or 
Reflex  (reference  8)  oscillators  which  can  be  easily  tuned  over  wide  bands.  After  adequate 
padding  these  sources  may  deliver  only  1-10  mw  to  the  slotted  line.  High-sensitivity 
detecting  and  indicating  methods  are  therefore  often  required.  Three  methods  are 
•common:  (1)  a  heterodyne  receiver  can  be  used;  (2)  a  simpler  method  is  to  100  per  cent 
modulate  the  source  by  a  square  wave  (to  avoid  frequency  modulation),  then  amplify 
and  rectify  the  detector  output  in  a  high-gain  amplifier-detector;  (3)  the  simplest  method 
is  to  use  a  sensitive  galvanometer  or  microammeter  driven  directly  by  a  crystal  detector. 
A  tunable  transformer  can  be  included  in  the  probe-detector  mechanism  to  give  the  high- 
est sensitivity  allowed  by  a  given  probe  insertion;  however,  any  variation  in  crystal  or 
bolometer  r-f  impedance  with  standing-wave  voltage  variation  may  cause  a  change  in  the 
tuning  and  introduce  a  significant  error.  Since  they  are  15-20  db  less  sensitive  than  crys- 
tals, bolometer  detectors  are  more  satisfactory  on  high-power  sources  (magnetron  test 
equipment,  reference  11)  for  which  high-sensitivity  methods  are  not  needed.  For  ac- 
curate measurements,  stabilized  power  supplies  must  be  employed,  and  spurious  pick-up 
in  microwave  oscillators,  high-gam  amplifiers,  and  detector  output  leads  must  be  avoided 
by  complete  shielding. 

Design  Requirements.  The  slot  should  be  as  narrow  as  possible,  of  uniform  width, 
accurately  centered  in  rectangular  wave  guide  and  parallel  to  the  axis  in  the  coaxial  line, 
and  long  enough  to  allow  a  probe  movement  of  at  least  1/2  wavelength.  To  prevent  radia- 
tion the  slot  can  be  covered  (by  a  contacting  shoe  or  a  shorting  trap)  for  at  least  1/4  wave- 
length on  each  side  of  the  probe.  To  avoid  serious  errors  caused  by  small  transverse  varia- 
tions of  the  probe  in  the  slot  the  probe's  outer  conductor  is  sometimes  imbedded  in  the 
•center  of  a  metal  slug  at  least  i/2  wavelength  long,  the  bottom  of  which  is  flush  with  the 
anside  of  the  outer  conductor  or  wave-guide  wall,  and  the  sides  of  which  definitely  wipe  (or 
definitely  dear  with  very  small  gaps)  the  sides  of  the  slot.  In  coaxial  slotted  lines  the  con- 
ductors must  be  accurately  coaxial.  The  center  conductor  can  be  rigidly  supported  and 
oentered  at  the  source  end,  but  at  the  load  end  it  should  be  supported  (if  necessary)  by  a 


IMPEDANCE  MEASUREMENTS  11-73 

very  thin  washer  of  low-dielectric-constant  material;  the  residual  reflection  (pure  shunt 
susceptance)  can  be  calibrated  and  allowed  for  subsequently.    An  assortment  of  low-reflec- 

itL  ^  T^f"8'  JaCkS'  Plug8'  ada*>ters>  may  ^  required  for  connection  to  the  load, 
although  greatest  accuracy  can  be  attained  with  as  few  attachments  as  possible.  Wave- 
guide slotted  hnes  are  preferably  terminated  with  a  well-surfaced  plane  flange,  to  which 
adapters  may  be  attached,  or  with  a  standard  choke  on  one  end  and  choke-cover  flange  on 
the  other  so  that  either  type  of  connection  is  available  by  reversal  of  the  slotted  section. 

Ihe  most  serious  requirement  is  that  the  bottom  of  the  probe  shall  travel  accurately 
parallel  to  the  axis  of  the  transmission  line.  The  probe  carriage  may  travel  on  the  outside 
of  the  line  itself,  or  on  ways  rigidly  attached  and  made  accurately  parallel  with  the  axis. 
It  is  generally  possible  by  accurate  machining  or  electroforming  techniques  (reference  12) 
to  maintain  parallelism  within  ±0.0002  in.  for  at  least  i/2  wavelength  of  travel. 

^  Accuracy.  On  standing-wave  ratios  above  about  2,  inaccurate  assumptions  (or  calibra- 
tions) for  the  law  of  the  detector  and  the  amplifying-indicating  system,  and  meter-reading 
errors,  can  easily  limit  the  measurement  accuracy  to  10  per  cent;  by  using  a  calibrated 
r-f  attenuator,  or  preferably  a  heterodyne  receiver,  or  by  deducing  the  voltage  standing- 
wave  ratio  from  measurements  confined  to  the  region  of  the  voltage  minimum  (reference 
3) ,  the  accuracy  can  be  improved  to  nearly  that  for  low  standing-wave  ratios.  Below  ratios 
of  2,  percentage  accuracy  is  limited  primarily  by:  (1)  change  in  characteristic  impedance 
and  end  reflections  introduced  by  the  slot;  (2)  reflections  introduced  by  load  connectors 
(flanges,  jacks,  tapers,  or  adapters) ,  unless  these  are  to  be  considered  a  part  of  the  unknown 
impedance;  (3)  reflections  from  the  probe;  and  (4)  non-parallelism  of  probe' travel.  The 
effect  of  (1)  can  be  deduced  by  measuring  an  impedance  through  two  Hnes  of  the  same 
dimensions,  one  with  and  the  other  without  a  slot.  Sometimes  errors  from  (2)  can  be 
reduced  by  determining  an  equivalent  shunt  susceptance  (reference  16),  but  this  is  im- 
practical when  the  connector  introduces  multiple  discontinuities.  It  is  possible,  but  not 
very  easy,  to  measure  and  calculate  errors  from  (3)  (reference  13);  this  effect  can  be 
reduced  by  maintaming  a  matched  impedance  looking  toward  the  source  and  eliminated 
by  withdrawing  the  probe  until  there  is  no  significant  change  in  power  delivered  to  the 
load  (as  monitored  by  another  detector);  but  a  compromise  must  usually  be  effected 
between  this  error  and  the  error  from  (4)  since  (4)  is  reduced  by  larger  probe  insertions. 

Experience  has  shown  that  carefully  constructed  coaxial  standing-wave  detectors  can 
be  used  with  an  accuracy  of  about  5  per  cent  and  the  accuracy  for  wave-guide  detectors  is 
usually  about  2  per  cent. 

DIRECTIONAL  COUPLERS.  Directional  couplers  (reference  14)  are  commonly  in- 
cluded as  a  permanent  section  of  the  coaxial  or  wave-guide  transmission  line  in  microwave 
systems  for  monitoring  incident  and  reflected  power  (reference  15) .  In  such  applications 
they  (1)  introduce  negligible  reflection  in  the  transmission  line  (S  less  than  1.05),  (2)  create 
almost  no  high-power  arcing  problem,  and  (3)  contain  no  moving  parts.  Compared  to 
standing-wave  detectors,  directional  couplers  can  be  used  for  routine  measurements  of  re- 
flections with  somewhat  less  accuracy  by  standard  techniques,  and  with  much  higher  ac- 
curacy by  special  techniques;  they  can  be  more  conveniently  used  (1)  to  tune  a  load  impe- 
dance to  match  the  line,  (2)  to  tune  a  source  impedance  to  match  the  line,  and  (3)  to  cor- 
rect for  changes  in  interaction  loss  between  the  source  and  load  impedance. 

Power  levels  proportional  to  the  power  in  the  reflected  and  incident  waves  can  be  meas- 
ured at  the  input  and  output  ends  respectively  of  an  auxiliary  transmission  line  coupled 
to  the  main  line  by  means  of  two  equal-sized  probes,  loops,  or  orifices  separated  by  approxi- 
mately 1/4  transmission-line  wavelength  (reference  14).  Illustrated  in  Fig.  3,  this  instru- 
ment comprises  a  directional  coupler  in  its  simplest  form.  A  directional  coupler  can  thus 
be  used  to  measure  reflection-coefficient 
amplitude,  which  is  the  ratio  of  the  square 

root  of  the  two  measured  power  levels,    source  i is^^t ~^r~7~ 

Bolometer  power  detectors  are  convenient  I  .,^ *a*t^I^'u  m^i-^ 

for  high-power  sources,  and  on  low-power 


sources  the  same  sensitive  detecting  and  |    |  "     AwdBary  fe»« 

indicating   methods  previously  described  R|^|d 

are  applicable.     It  is  preferable  to  switch          wave  output 

the  same  detector-indicator  alternately  be-          jr^  3     Directional  Coupler  in  Waveguide 

tween  incident  and  reflected  wave  outputs 

(alternately  terminating  the  other  output)  so  that  only  a  ratio  calibration  is  required. 

Low-reflection  wave-guide  and  coaxial  switches  are  convenient  for  this  purpose,  or  special 

arrangements  can  be  employed  to  transmit  both  waves  to  the  same  output.     Detector 

law  uncertainties  can  be  eliminated  by  using  a  calibrated  r-f  attenuator  to  equalize  incident 

and  reflected  powers.    The  attenuator  method  is  best  used  for  measuring  nearly  matched 

impedances,  since,  for  example,  a  change  in  standing-wave  ratio  from  1.01  to  1.02  changes 


11-74 


ELECTRICAL  MEASUREMENTS 


Oscillator 

Attenuator 

- 

Tunable 
matching 
network 

I                       t 

3  

I 

Y   Snorting 
piston 

A.  Tuning  source  impedance 
to  match  tne  transmission  line 


I                               i 

1 

_deal 
ource 

Source 

^*&mmWb^ 

s 

'                     \^      l  $fa 

Adjustable 
attenuator 

n 

Detector 
and 
Indicator 

B,  Simulation  of  perfectly  stable,  matched  source 


the  reflected  wave  output  power  by  6  db  whereas  one  from  2.02  to  2.04  changes  the  output 
power  by  only  0.12  db.  .         .      ,  , 

A  directional  coupler  can  thus  be  used  as  a  sensitive  indicator  for  tuning  a  load  to  match 
the  transmission  line,  and,  because  reflection  coefficient  is  continuously  monitored,  it  is 
especially  convenient.  ,  ..  . 

In  Fig.  4  a  directional  coupler  is  used  for  matching  a  source  impedance  to  the  line.  As  a 
shorting  piston  (or  preferably  a  reflector  with  a  constant  K  of  30  to  50  per  cent)  is  moved 
back  and  forth  in  the  line,  the  tuning  controls  are  adjusted  until  the  incident  wave  output 
remains  constant.  The  oscillator  must  be  sufficiently  masked  or  decoupled  to  prevent 

changes  in  its  frequency  and 
efficiency  as  the  load  impedance 
•  varies. 

Without  tuning  for  it,  the 
condition  prevailing  for  a 
matched  source  impedance 
(Fig.  7)  can  be  simulated  by 
monitoring  and  keeping  the  in- 
cident power  constant  by  means 
of  a  variable  attenuator  ahead 
of  the  coupler,  Fig.  4.  The 
variation  of  power  in  the  load 
as  its  impedance  changes  is  re- 
duced to  that  of  reflection  loss 
only.  Power-time  instability  in 
the  oscillator  can  also  be  cor- 
rected for  by  this  method. 

Design  Requirements.  Di- 
rectional couplers  for  imped- 
ance measurement  should  not 
have  losses  much  below  20  db 
(see  p.  11-72)  to  avoid  serious 
loading  by  the  coupling  holes 
and  interaction  between  them. 
By  careful  machining  or  electro- 
forming  techniques  (reference 
12),  inside  dimensions  and  the 
wall  thickness  of  the  main  line 
must  be  made  precisely  uniform 
over  the  longitudinal  region  oc- 
cupied by  the  coupling  slits  or 
holes,  which  must  be  con- 
structed with  exacting  tolerances.  Transverse  slits  have  been  used  both  in  coaxial  lines 
and  in  wave  guides  (usually  in  the  wide  side),  but  for  wave-guide  couplers  round  holes  in 
the  narrow  side  are  usually  better  since  they  give  higher  losses  which  can  be  more  accurately 
controlled  by  standard  precision  boring  techniques.  In  coaxial  lines  the  center  conductor 
must  be  accurately  centered;  reflection  from  any  support  at  the  load  end  cannot  be  cali- 
brated out  of  the  measurements  which  ordinarily  do  not  contain  phase  information.  Simi- 
larly, careful  attention  must  be  given  to  designing  refleetionless  fittings  and  connectors. 
The  most  serious  requirement  for  accurate  measurements  is  that  the  directional  coupler 
be  designed  for  the  frequency  band  in  which  it  is  to  be  used.  Its  directional  coupling 
action  (directivity)  is  frequency  sensitive.  As  Xg,  the  transmission-line  wavelength  (not 
to  be  confused  with  free-space  wavelength)  varies,  the  effect  of  a  fixed  hole  spacing  is  to 
add  a  spurious  vector  voltage  to  the  true  reflected  wave  output  in  the  auxiliary  line.  The 
ratio  of  this  spurious  voltage  to  the  incident  wave  voltage  (also  in  the  auxiliary  line)  shall 
be  designated  "unbalance  reflection  coefficient,"  Ku.  In  the  worst  case,  a  measured  value 
of  reflection  coefficient  will  be  in  error  by  ±KU.  For  a  simple  two-hole  coupler  in  wave 
guide,  Ku  is  zero  at  a  Xg  approximately  1  per  cent  greater  than  four  times  the  hole  spacing, 
but  for  a  ±0.6  per  cent,  ±1.3  per  cent,  and  ±3.2  per  cent  change  in  Xg  the  values  for  Ku 
are  approximately  0.01,  0.02,  and  0.05  respectively.  The  band  width  can  be  greatly 
increased  by  increasing  the  total  number  of  holes,  and  the  simplest  pattern  is  to  separate 
equal-sized  pairs  of  holes  (spaced  1/4  the  mid-wavelength)  by  1/2  mid-wavelength.  By 
this  method  Ku  can  be  kept  lower  than  0.015  over  an  8  per  cent  wavelength  band  in  a  four- 
hole  wave-guide  coupler  and  over  a  20  per  cent  band  for  an  eight-hole  wave-guide  coupler 
(reference  26). 


C.  Precision  measurement  of 

reflection  coefficient 
FIG.  4.     Directional  Coupler  Applications 


IMPEDANCE  MEASUREMENTS 


11-75 


•anch 


H  branch 


>L  Hybrid  junction  in  waveguide 


Another  spurious  vector,  adding  to  the  true  reflected  wave  output,  comes  from  partial 
reflection  of  the  incident  wave  from  an  imperfect  termination  of  the  auxiliary  line.  Where 
it  is  practical  to  position  this  termination  for  first  a  minimum  then  a  maximum  reading,  it 
is  possible  to  nearly  eliminate  its  error  by  averaging  the  two  readings. 

Accuracy.  In  the  laboratory  where  there  is  usually  no  serious  limitation  on  length, 
it  is  practical  to  use  four-  or  eight-hole  couplers  containing  broad-band  terminations  with 
VSWR  less  than  1.05  in  coaxial  and  1.02  in  wave  guide.  Under  these  conditions,  routine 
measurements  can  be  made  with  a  VSWR  accuracy  of  about  10  per  cent  in  coaxial  and 
5  per  cent  in  wave  guide,  since  detecting-indicating  systems  can  be  made  to  contribute 
small  percentage  errors.  In  microwave  systems  directional  couplers  are  often  limited  in 
size  so  that  the  realizable  accuracy  is  usually  lower. 

Figure  4  shows  a  method  wherein  the  auxiliary  line  termination  is  tuned  to  cancel  the 
Ku  of  the  holes,  giving  the  effect  of  a  perfect  coupler  and  termination.  As  a  mismatch  of 
constant  K  is  moved  back  and  forth  in  the  main  line,  the  termination  tuning  is  adjusted 
until  the  reflected  wave  output  remains  constant  (not  zero) .  Alternatively,  if  a  termination 
of  K  —  0  is  used  the  output  can  be  directly  tuned  to  zero.  Loads  (including  their  con- 
nectors) connected  to  the  directional  coupler  can  then  be  measured  with  an  accuracy  of 
about  0.1  to  0.2  per  cent.  Accuracy  is  limited  primarily  by  the  detecting-indicating 
equipment. 

HYBRID  JUNCTIONS.  Hybrid  junctions  (reference  18)  can  be  used  in  many  of  the 
applications  described  for  directional  couplers,  compared  to  which  they  may  possess  certain 
advantages  in  sensitivity,  size,  design  simplicity,  and 
band  width.  It  is  possible  to  make  hybrid  junctions 
in  either  wave-guide  or  coaxial  lines.  However,  a  de- 
sign which  will  give  accurate  measurements  over  a 
wide  bandwidth  does  not  yet  exist  for  coaxial  lines. 
Descriptions  and  examples  in  the  following  discussion 
are  applied  specifically  to  wave-guide  hybrid  junctions 
for  which  satisfactory  designs  are  commonplace. 

A  type  of  hybrid  junction  commonly  used  for  imped- 
ance measurement  is  formed  by  joining  an  J?-plane  T 
(off  the  narrow  side)  and  an  ^-plane  T  (off  the  wide 
side)  to  a  straight  section  of  rectangular  wave  guide 
(reference  12) .  Projections  of  the  central  axes  of  both 
T's  meet  at  a  common  point  on  the  axis  of  the  main 
guide,  Fig.  5.  This  geometrical  symmetry  makes  the 
E  and  H  branches  conjugate.  When  the  main 
branches  are  terminated  by  equal  impedances  there  is 
no  transmission  between  E  and  H  branches.  Hybrid 
junctions  can  therefore  be  used  to  tune  a  load  on  one 
main  branch  (test  branch)  to  match  that  on  the  other 
main  branch  (reference  branch) ;  the  degree  to  which, 
the  tuned  load  matches  the  characteristic  impedance 
of  the  wave  guide  depends  upon  (1)  the  intrinsic  bal- 
ance in  the  junction  and  (2)  the  reflection  coefficient 
of  the  reference  termination.  The  directional  coupler 
has  corresponding  limitations;  in  fact,  both  devices 
separate  out  direct  and  reflected  waves  by  a  cancella- 
tion process.  However,  it  is  sometimes  more  helpful  to  visualize  the  hybrid  junction  as 
the  microwave  equivalent  of  a  low-frequency  hybrid  transformer  (reference  18)  with  the 
added  complication  that  the  input  impedances  at  the  junctions,  unless  matched,  are 
transformed  to  new  values  along  the  transmission-line  branches. 

Looking  toward  the  junction  the  impedances  are  inherently  unmatched  to  the  branch 
transmission  lines;  on  well-constructed  junctions  the  VSWR's  looking  alternately  into 
E,  H,  test,  and  references  branches  are  approximately  2,  3,  1.3,  and  1.3  respectively  when 
the  other  three  branch  lines  are  terminated  in  ZQ. 

By  methods  similar  to  those  described  for  the  directional  coupler,  hybrid  junctions  can 
also  be  used  for  routine  and  precise  measurements  of  reflection  coefficient.  Figure  5 
illustrates  the  circuit  configuration.  The  ^-branch  output  (detector  input)  is  proportional 
to  the  vector  sum  of  the  reflected  waves  in  the  test  and  reference  branches,  and  a  wave 
created  by  junction  unbalance.  When  the  latter  two  are  negligible,  the  detector  input  is 
proportional  only  to  the  power  reflected  from  the  unknown  impedance;  reflection  coefficient 
can  be  deduced  from  the  ratio  of  this  power  to  the  power  incident  upon  the  unknown.  For 
sensitivity  calibration  a  power  level  proportional  to  the  incident  wave  (in  the  test  branch) 


B.  Measurement  of  reflection  coefficient 
FIG.  5.     Hybrid  Junction 


11-76  ELECTRICAL  MEASUREMENTS 

is  most  conveniently  applied  to  the  detector  by  substituting  a  shorting  plunger  for  the 
unknown.  .  7  a  ,  , 

As  a  consequence  of  its  reflection  coefficient,  kt,  the  test  branch  carries  a  total  reflected 
wave,  Vr,  which  is  not  uniquely  proportional  to  the  product  of  the  reference  incident  wave 
Vi  and  the  unknown  reflection  coefficient  kx.  In  fact,  Vr  =  Vikx/(l  -  kxkt),  which  is  a 
complex  quantity.  Evidently  the  amplitude  ratio  VT/Vi  as  measured  on  the  hybrid 
junction  is  not  the  true  reflection  coeflicient  Kx\  therefore  the  maximum  error  in  routine 
measurements  created  by  Kt  (assuming  that  the  effect  is  not  canceled  by  purposely 
phasing  the  unknown  along  the  line)  is  1/(1  -  K*Kt).  For  example,  a  measurement 
which  gave  Kx  =  9.1  per  cent  (SWR  =  1.2)  may  be  in  error  by  a  maximum  factor  of  1.01 
since  Kt  =  13  per  cent.  The  error  increases  at  higher  values  of  Kx,  but  nearer  zero  the 
error  from  this  source  will  be  negligible  (reference  27) . 

When  a  short  circuit  is  substituted  for  the  unknown  for  sensitivity  calibration,  Kx  =  1. 
As  tbe  position  of  the  short  circuit  is  varied  the  power  in  the  detector  will  deviate  from  the 
reference  value  by  maximum  factors  of  1/(1  4-  Kt)  to  1/(1  -  Kt}.  From  an  observation 
of  the  maximum  and  minimum  detector  outputs  it  is  possible  to  deduce  that  reference 
detector  output  which  is  proportional  to  the  reference  incident  power  in  the  test  branch, 
However,  the  impedance  presented  to  the  source  will  be  seriously  altered  by  the  short 
circuit  in  the  test  branch,  so  the  above  factors  can  be  applied  exactly  only  when  the  oscil- 
lator is  well  masked  and  the  source  impedance  is  well  matched  to  the  line;  alternatively 
the  source's  incident  power  can  be  monitored  by  a  directional  coupler  and  kept  constant, 
Fig.  4.  The  detector  must  also  present  a  well-matched,  ZQ,  load.  An  alternative  method  of 
calibration  which  reduces  the  interaction  effects  occasioned  by  introducing  a  shorting 
piston  is  to  substitute  a  known  impedance  (calibrated  by  other  means)  of  medium  or 
low  K.  In  this  case,  the  net  measurement  accuracy  can  be  no  better  than  that  of  the 
calibrated  impedance. 

The  above  interaction  effects  can  be  materially  reduced  or  even  eliminated  by  introduc- 
ing properly  positioned  susceptances  (references  5  and  17)  (matching  posts,  windows,  or 
dielectric  blocks)  which  cancel  the  junction's  discontinuity  reactances  and  transform  the 
impedance  of  the  H  and  E  branches  so  that  each  branch  is  matched  when  the  other  three 
are  terminated  with  matched  loads.  Under  this  condition,  the  main  branches  are  also 
conjugate,  and  power  sent  into  any  one  branch  divides  equally  between  the  two  non- 
conjugate  branches  (reference  IS).  Such  an  "ideal"  hybrid  junction  can  be  used  in  all 
those  applications  described  for  directional  couplers,  except  monitor,  without  disturbing 
the  transmission  line,  with  as  good  accuracy  and  higher  sensitivity.  For  example,  high- 
er low-impedance  mismatches  can  be  measured  accurately,  a  load  can  be  tuned,  and 
a  source  impedance  can  be  tuned  to  ZQ  by  monitoring  and  tuning  for  constancy  the 
incident  wave  output  on  one  main  branch  while  moving  a  shorting  piston  in  the  other  main 
branch. 

The  above  impedances  may  be  matched  over  only  a  relatively  narrow  band,  unless  the 
posts  or  windows  are  put  right  in  the  junction.  However,  the  unbalance  reflection  coeffi- 
cient may  be  seriously  increased  by  this  method  of  impedance  matching.  Completely 
satisfactory  wide-band  solutions  have  not  yet  been  found.  Matching  networks  are  not 
essential  for  many  measurement  requirements. 

Accuracy.  Accuracy  of  routine  measurements  is  primarily  limited  by  (1)  interaction 
between  unknown  and  test  branch  impedances,  (2)  reflection  coefficient  of  the  reference 
termination,  and  (3)  junction  unbalance  reflection  coefficient.  For  low  mismatches  (K 
less  than  5  per  cent)  the  first  error  is  negligible,  and  under  this  condition  the  same  tech- 
niques described  for  directional  couplers  can  be  used  to  (a)  cancel  the  second  error  by 
positioning  the  reference  termination  and  (6)  cancel  both  second  and  third  errors  by 
timing  the  reference  terminations  for  precision  measurements.  On  hybrid  junctions 
constructed  by  precision  electroforming  techniques  (reference  12)  the  unbalance  reflection 
coefficient  can  be  kept  less  than  0.005  over  the  entire  wave-guide  pass  band.  Therefore 
on  low  mismatches  the  accuracy  of  routine  measurements  can  be  as  high  as  or  higher  than 
on  well-constructed  directional  couplers  with  the  added  advantage  that  lower-sensitivity, 
more  stable  detecting  and  indicating  equipment  can  be  used. 

Compared  to  directional  couplers,  hybrid  junctions  have  the  following  advantages: 
(1)  they  are  simpler  to  design  for  a  low  unbalance  reflection  coefficient  over  a  wide  band 
(geometrical  symmetry  is  the  only  requirement);  (2)  they  are  smaller;  and  (3)  they  are 
more  sensitive  (the  reflected  wave  power  in  the  detector  is  at  least  10  db  higher  than  that 
for  a  20-db  directional  coupler) .  However,  (1)  they  cannot  be  incorporated  in  transmission 
lines  (with  negligible  reaction)  to  monitor  reflection  coefficient;  (2)  interaction  between 
"unknown"  and  test  branch  impedances  may  introduce  a  significant  error  in  measuring 
reflection  coefficients  higher  than  about  5  per  cent  (S  =  1.1) ;  and  (3)  impedance  interaction 
effects  make  it  more  difficult  to  calibrate  sensitivity  without  introducing  a  calibration  error. 


ABSOLUTE  POWER  MEASUREMENTS 


11-77 


33.  ABSOLUTE  POWER  MEASUREMENTS 


Brrdge  circuit 
connection 


Quarterwa 
stub  supp 


R-f  input 


Dielectric- 
;      r^  bypass 


\  T 
r 


Thermistor  bead 
•obe  aoienna 


.  la  waveguide 


R-f  Input 


TWVOT       l        +^       T  n  on          BY   ^OLOMETRIC    METHODS.      Microwave 

powers  less  than  10-20  mw  are  measured  by  bolometric  methods  which  are,  so  far,  the 
only  ones  available  for  accurate  measurements  in  this  frequency  and  power  range.  A 
transmission  line  carrying  the  unknown  power 
is  terminated  in  a  bolometer  detector,  the  only 
absorbing  element  of  which  is  a  thermally  sensi- 
tive resistor.  Its  d-c  resistance  changes  when 
r-f  power  is  dissipated  in  it,  and  ideally  the 
change  is  independent  of  the  frequency  of  exci- 
tation. The  resistance  change  can  therefore  be 
related  to  r-f  power  by  a  low-frequency  or  d-c 
calibration.  Alternatively  the  resistance  can  R-f  «nj 
be  biased  to  a  given  value  and  kept  constant 
when  r-f  power  is  applied  by  removing  an  equal 
and  measured  quantity  of  d-c  or  low-frequency 
power. 

A  bolometer  detector  (Fig.  6  is  typical) 
comprises  a  thermal  resistor  and  a  reactance 
network  for  matching  it  to  the  transmission 
line.  Sensitive  thermal  resistors  are  small, 
essentially  "lumped,"  elements.  Two  types 
are  commonly  employed:  thin,  short,  filament 
wires  (reference  19),  and  bead  thermistors, 
the  thermal  element  of  which  is  a  tiny  bead 
made  up  of  a  mixture  of  metallic  oxides  (ref- 
erence 8)  (see  Table  2).  The  r-f  resistance 
of  the  thermal  resistor  may  be  different  from 
d-c  resistance,  but,  in  order  to  compare  r-f 
power  directly  against  d-c  (or  low-frequency) 
standards,  (1)  its  d-c  resistance  change  must 
be  dependent  only  upon  incremental  heating 
power  (when  external  temperature  is  constant), 
(2)  the  heat  distribution  from  r-f  power  must 
be  equivalent  to  that  generated  by  a  uni- 
formly distributed  d-c  current,  and  (3)  all 
the  r-f  power  absorbed  by  the  detector  must  be 
dissipated  in  the  thermal  resistor. 

Requirement  1  is  met  on  all  thermal  resistors  when  space-integrated  values  of  resistance 
and  heat  are  specified.  However,  the  time  constant  of  the  thermal  resistor  may  be  of 
such  a  value  that,  in  two  extreme  cases,  resistance  changes  either  (a)  exactly  follow  or 
(6)  completely  fail  to  follow  the  modulation  envelope  of  the  r-f  power  (when  amplitude 
modulated) .  In  either  case  the  average  resistance  change  is  proportional  to  average  power, 
but  in  (a)  envelope  peak  power  can  be  measured-  As  sinusoidal  modulation  frequency  is 
increased  from  zero  (all  other  parameters  remaining  constant)  the  resistance  modulation 
will  decrease  to  half  its  maximum  value  at  36  cps  and  at  450  cps,  respectively,  for  the  ther- 
mistors and  platinum  wire  listed  in  Table  2. 

Table  2.     Sensitive  Thermal  Resistors — Typical  Characteristics 


Dielectric 
r-f  bypass 


I?.  In  coaxial 

FIG.  6.     Pre-tuned,  Wide-band  Bolometer  De- 
tectors 


Platinum  Wire 

Thermistor  Bead 

Temperature 

25°  C 
200  ohms 
15.3mw 
+4.5  ohms/mw 

—  0.0  5mw/deg  cent 
350  Atsec 
11  ma 
0.118  X  0.00006  in. 

25°  C 

200  ohms 
10  mw 
—  29  ohms/mw 

Lead' 
Bead, 

25°  C 
125  ohms 
I3.5mw 
—  1  4  ohms/mw 

—  0.1  mw/deg  ce 
2500  fjsec 
200  ma 
wares,  0.001  in.  d 
0.020  X  0.0  10  in 

25°  C 

50  ohms 
23.5  mw 
—  4.8  ohms/mw 

at 

iameter 
t. 

Resistance     .                        ... 

Resistance-power  coefficient  .  . 
Power-temperature        coeffi- 
cient * 

Time  constant  "f  

Safe  rnaxiTTiuTn  ciirrpnt       -  - 

Dimensions  

Tor  constant  resistance;  also  the  ratio  of  power  to  temperature  changes  which  produce  the  same 
resistance  changes. 

f  For  67  per  cent  of  ultimate  resistance  change. 


11-78 


ELECTRICAL  MEASUREMENTS 


SJS     0.2 

£  = 


Source  VSWR> 


Requirement  2  is  met  on  filament  wires  by  (a)  designing  a  matching  configuration  which 
places  the  r-f  current  maximum  at  the  midpoint  of  the  wire,  and  (6)  by  limiting  the  upper 
frequency  to  that  corresponding  to  a  free-space  wavelength  of  about  8  times  the  length 
of  the  wire  (reference  7).  Platinum  wires  have  given  satisfactory  results  up  to  10,000 
megacycles.  Bead  thermistors,  which  are  more  nearly  ideal  "lumped"  elements,  have 
been  used  up  to  25,000  megacycles,  where  they  have  shown  discrepancies  less  than  5  per 

cent  when  checked  against  calo- 
0-6 {          '         ' '         '         ' '         ' '     rimeter  standards. 

Requirement  3  must  be  met  by 
transforming  the  a-c  resistance 
component  to  the  transmission 
line's  characteristic  impedance 
with  a  matching  network  in  which 
considerable  care  is  taken  to  elimi- 
nate extraneous  circuit  losses  from 
sliding  contacts,  joints,  and  con- 
ductors. This  problem  is  simpli- 
fied by  (a)  biasing  the  thermal  re- 
sistor to  a  reasonable  resistance 
(50-200  ohms)  and  by  (6)  design- 
ing a  low-Q  matching  configura- 
tion, which  tends  to  eliminate  high 
current  concentrations  (standing 
waves)  in  the  detector. 

The  power  absorbed  (and  there- 
fore measured)  by  a  detector  load 
is  a  function  of  both  source  and 
load  impedances,  and,  depending 
upon  relative  phase  angles  (or  line 
length),  varies  between  the  limits 
shown  in  Fig.  7  (reference  20).  To 
eliminate  these  uncertainties  in 
measurement  and  interpretation  of 
power  quantities  it  is  evidently  de- 
sirable to  use  a  detector  that  pre- 
sents a  nearly  matched  impedance. 
Wide-band,  pretuned  detectors,  in 
addition  to  being  especially  con- 
venient to  use,  inherently  meet  the 
low-£  requirement.  Table  3  lists 
the  band  widths  that  have  been 
realized  for  typical  pretuned  bo- 
lometer detectors  using  thermi- 
stors. 

Since  the  r-f  impedance  varies  with  it,  the  thermal  resistor's  d-c  resistance  must  be 
kept  nearly  constant  by  varying  (with  temperature)  the  biasing  power  by  methods  peculiar 
to  the  bridge  measuring  circuit. 

Table   3.     Bandwidth-Impedance    Characteristics    of   Typical   Pretuned   Thermistor 

Detectors 


-0.6 


For  the  above  VSWRs,  line  lengths  are  chosen  so  that 
Pj  =  least  power  and  Pnt=most  power  in  the  load. 
Available  power  Ffc  could  be  delivered  if  conjugate 
impedence- matching  transformers  were  used  in  the  line. 

FIG.  7.    Power  Delivered  by  a  Mismatched  Source  to  a  Mis- 
matched Load 


Frequency  Band 
in  Megacycles 

Maximum 
VSWR 

Transmission  Line 

Thermistor  Operating 
Resistance  in  Ohms 

5  to  600  . 

2 

Coaxial 

Ch             '  f    7    f  r 

700  to  1500  

4 

Coaxial 

100 

2900  ±  17  2% 

4 

inn 

3700  ±8.1%     

4 

Coaxial 

inn 

4500  ±111% 

4 

1  T^ 

4100  ±  12.2% 

2 

oaxia 

7c 

4600  ±8.2%  

1    l 

2  in.  X  I  in.  OD 

7r 

9.050  ±6.1%    

I  4 

2  in.  X  1  in.  OD 

1  'JC 

24,0€0  ±  4.2%  

1   4 

1  1/4  in.  X  5/8  in.  OD 

1/2  in.  X  1/4  in.  OD 

ABSOLUTE  POWER  MEASUREMENTS 


11-79 


MTHiammeter 


'Microammeier 


Biasing  and  power-indicating  circuits  are  designed  to  accom- 
as  those  shown  in  Table  2.     Resistance  changes  are  most 

f  •    ,      -,-   .       i  -     -  ~ ge  c*rcufts  such  as  Fig.  S  which  illustrates  the  simplest  method 

°*  }hlrdiff^S  f8??  ?°Wel  a ud  measuring  ^  P^er.  Power  can  be  directly  measured 
as  the  difference  of  the  two  d-c  biasing  powers  required  for  reference  resistance,  one  before 
d^dva^l  r'f  P°wer  ls  fPPHed.  However,  this  simple  method  contains  three  important 
disadvantages.  (1)  since  the  r-f  power  may  be  a  small  difference  of  two  relatively  large 
d-c  powers,  small  meter-reading  errors  may  introduce 
large  errors  in  the  difference  power  measurement;  (2) 
r-f  power  is  not  continuously  indicated;  (3)  the  unbal- 
ance caused  by  external  temperature  changes  subse- 
quent to  initial  balance  conditions  are  indistinguishable  />  ^_TA_IX  Bolometer 
from  r-f  power  changes.  <L  X^ZL,  /deiSs 

Difference  errors,  (1) ,  can  be  reduced  by  interposing  a       T  X>— -3    * JjJS"1 

resistance  network  between  the  bridge  and  a  current  or    ^       e     e 

voltage  source  which  is  kept  constant  at  a  precisely  mea-    Fl<J'  8'  B^^^S™**  f°r 
surable  single  value.    The  network  must  contain  accu-  Bolometer  Detectors 

rately  known  resistance  elements,  one  or  more  of  which  are  switched  in  or  out  in  small  and 

™^ni?tepS<    Biasing  Powers  are  then  deduced  from  settings  on  the  switch. 

Within  the  accuracy  limitations  imposed  by  (3),  unbalance  current  can  be  used  as  a 
continuous  indication  of  relative  r-f  power  level;  absolute  power  can  be  deduced  from  a 
calibration  of  the  unbalance  sensitivity.  This  sensitivity  calibration  will  be  an  uncritical 
but  not  negligible  function  of  temperature  unless  the  d-c  voltage  across  the  bridge  can 
be  kept  constant  (this  can  be  made  possible  by  using  variable  low-frequency  biasing  power 
to  balance  the  bridge  initially) .  Resistance  unbalance  must  be  limited  to  a  maximum  value 
dictated  by  the  detector's  r-f  matching  requirements. 

The^lowest  power  that  can  be  accurately  measured  is  limited  by  (3)  and  the  temperature 
coefficient  of  the  thermal  resistor.  For  example,  from  Table  2,  a  subsequent  temperature 
change  as  small  as  ±0.1  deg  cent  would  cause  a  3-db  error  in  measuring  powers  of  2.5-5  juw 
and  0.1  db  in  measuring  powers  of  0.25-0.5  mw.  These  errors  can  be  materially  reduced 
by  (a)  using^thermal  insulation,  (6)  using  large  metal  masses  to  limit  the  rate  of  temperature 
change,  (c)  interposing  between  the  source  and  the  detector  an  essentially  instantly  oper- 
able cut-out  switch,  reactive  gate,  or  attenuator  which  can  then  be  used  to  remove  r-f 
power  rapidly  to  check  the  initial  balance,  and  (d)  using  temperature-compensating 
bridges.  A  combination  of  all  the  above  techniques  is  usually  required  for  accurate  power 
measurements  in  the  range  of  5-100  /iw. 

A  practical  solution  to  the  disadvantages  of  the  simple  bridge  circuit  has  been  to  com- 
bine all  the  remedies  into  a  more  complex  circuit  such  as  Fig.  9.  The  thermal  resistor  in, 


Calibrating 
circuits 


Regulating  \\  Bridge-regulated 
bridge         \\      osoiLaior 


Compensating-^ 

thermal  resistor 


k— R-f  input 


Measuring 
therm aJ  resistor 


FIG.   9. 


Temperature-compensating,  Self-calibrating,  Direct-reading  Bridge  Circuit  for  Bolometer 

Detectors 


the  bolometer  detector  is  used  as  the  regulating  element  in  a  conventional  bridge-regulated 
oscillator.  The  oscillator  automatically  delivers  enough  power  to  the  thermal  resistor  to 
bias  its  resistance  very  close  to  the  bridge-balancing  value  (within  1  per  cent).  The  oscilla- 
tion power  level  will  therefore  vary  with  ambient  temperature  with  a  coefficient  determined 
by  the  temperature  coefficient  of  the  thermal  resistor.  From  Table  2,  this  coefficient 
would  be  —0.05  mw/deg  cent  for  the  platinum  wire  and  —0.1  mw/deg  cent  for  bead 
thermistors. 

The  output  of  the  low-frequency  oscillator  is  also  delivered  to  another  bridge  which 
contains  a  compensating  and  measuring  thermal  resistor  (never  excited  by  r-f  power)  of 


11-80  ELECTRICAL  MEASUREMENTS 

the  same  type  as  that  used  in  the  bolometer  detector.  When  RI  is  properly  adjusted  the 
power  received  by  the  compensating  resistor  varies  with  temperature  at  a  rate  equal  to 
that  required  by  this  resistor  to  be  biased  to  a  constant  resistance.  Both  thermal  resistors 
must  be  subjected  to  identical  external  temperature  variations.  Intimate  thermal  cou- 
pling can  best  be  accomplished  by  burying  the  compensating  thermal  resistor  in  the 
same  metal  mass  (preferably  large)  which  forms  the  outside  boundaries  of  the  bolometer 
detector. 

As  a  net  result  the  measuring  bridge  remains  balanced,  even  though  temperature 
changes,  until  r-f  or  incremental  d-c  power  is  applied  to  the  bolometer  detector.  The  d-c 
biasing  powers  are  kept  constant  so  that  the  unbalance  sensitivity  of  the  measuring  bridge 
does  not  vary  with  temperature.  The  overall  sensitivity  is  established  at  one  power  level 
by  using  the  d-c  calibrating  network  which  introduces  a  known  increment  of  d-c  heating 
power  in  the  bolometer  detector.  Other  unbalance  current  readings  can  be  related  to  the 
calibrated  point  from  a  knowledge  of  the  indicator  law.  Both  the  measuring  bridge  net- 
work and  the  intrinsic  characteristics  of  its  thermal  resistor  influence  the  law  of  unbalance 
current  versus  power  input.  In  general,  this  law  is  not  exactly  linear,  but  it  can  be  accu- 
rately calibrated  by  exciting  the  detector  with  a  convenient  low-frequency  source  whose 
relative  output  power  is  varied  in  known  ratios  by  a  low-frequency  attenuator. 

Accuracy.  It  is  usually  practical  to  limit  errors  from  r-f  sources  to  very  low  values, 
and  then  power-measurement  accuracy  is  primarily  governed  by  the  d-c  calibrating  and 
indicating  circuits.  The  circuit  of  Fig.  9  can  be  used  to  measure  full-scale  powers  with 
less  than  ±0.1  db  error,  half-scale  powers  within  ±0.2  db,  and  quarter-scale  powers  within 
±0.3  db.  On  an  indicating  meter  whose  resistance  is  equal  to  the  bridge  arms,  deflections 
of  200  and  50  microamperes  per  milliwatt  are  typical  for  thermistors  and  platinum  wires 
respectively.  Therefore  powers  below  1/4  mw  cannot  be  very  accurately  measured  unless 
meter-reading  errors  are  kept  low  by  using  sensitive  galvanometers  or  very  stable  indicator 
amplifiers;  it  is  then  possible  to  attain  accuracies  of  ±0.4  to  0,6  db  down  to  5-10  juw. 
However,  the  severe  temperature  drift  problems  must  be  solved  by  exacting  methods  as 
discussed  above. 

MEDIUM  AND  HIGH-POWER  MEASUREMENTS  BY  BOLOMETRIC  METHODS. 
Bolometer  detectors  containing  thermal  resistors  which  are  less  sensitive  and  require 
higher  biasing  powers  than  those  in  Table  2  (references  7,  8,  27)  can  be  used  to  measure 
powers  up  to  100-200  mw.  In  general  the  methods  are  the  same  as  those  outlined  above. 
Alternatively,  fixed  or  variable  r-f  attenuation  of  known  value  can  be  interposed  between 
the  source  and  a  low-power  bolometer  detector;  this  method  is  the  more  flexible  since 
power  levels  are  restricted  only  by  the  power-handling  capability  of  the  attenuator.  Since 
attenuators  capable  of  dissipating  power  up  to  several  hundred  watts  (up  to  1000  watts 
in  some  designs)  have  been  realized  in  practice,  the  power  range  inherent  in  this  method 
is  at  least  0-100  watts. 

The  attenuator's  attenuation  may  be  a  significant  function  of  frequency  so  that  a 
calibration  should  be  made  (or  obtained  from  the  known  frequency  characteristic)  at  the 
measurement  frequency.  Interaction  between  the  impedance  of  the  source  and  the  im- 
pedance at  the  attenuator's  input,  and  between  the  impedance  of  the  detector  and  the 
impedance  at  the  attenuator's  output,  creates  two  sources  of  uncertainty  in  power  measure- 
ment (see  Fig.  7).  These  impedance  interactions  are  preferably  reduced  or  eliminated 
by  an  attenuator  design  which  gives  nearly  unity  match  when  alternate  terminals  are 
terminated  in  unity  match. 

Errors  from  attenuator  calibration  instability  with  respect  to  time,  ambient  temperature, 
power  input  level,  and  normal  handling  can  usually  be  reduced  to  insignificance  by  (1) 
selection  of  an  appropriate  type  of  attenuator,  (2)  careful  attention  to  design  and  construc- 
tion details,  and  (3)  frequent  recalibration  when  necessary. 

The  most  stable  fixed  coaxial  attenuator  types  have  been  (1)  resistance-film  center 
conductor,  (2)  lumped-element  TT  or  T  pad,  and  (3)  lossy  dielectric.  The  first  two  types 
are  generally  stable  up  to  1  watt;  using  lossy  ceramics  and  heat-radiating  fins  type  (3) 
can  be  designed  to  handle  100-1000  watts  stably.  Lossy-dielectric  flexible  cables  have 
been  generally  unsatisfactory.  In  wave  guides,  resistance-film  (parallel  to  the  electric  field) 
types  are  used  up  to  1  watt,  and  lossy  ceramic  dielectric  types  up  to  100  watts.  In  many 
respects  the  most  satisfactory  fixed  attenuators  (in  either  coaxial  or  wave  guide)  have  been 
directional  couplers.  Attenuation  stability  is  unquestionably  high.  Attenuations  above 
10-15  db  give  the  best  impedance  characteristics.  Stringent  directivity  requirements  are 
not  necessary.  Power-handling  capability  is  limited  solely,  and  input  impedance  primarily, 
by  the  main  line  termination,  which  may  be  a  useful  load  or  a  dummy  load  of  the  lossy 
dielectric  type  (reference  15)  or,  for  low  powers,  a  resistance-film  termination.  The  output 
impedance  is  governed  primarily  by  the  low-power  termination  in  the  auxiliary  guide. 
It  is  practical  to  limit  input  and  output  impedance  mismatches  to  1.1  and  1.05  VSWR  in 


11-82  ELECTRICAL  MEASUREMENTS 

cent  higher  than  that  of  the  inlet  water  and  the  rate  of  flow  is  m  grams  per  second  then 
w  =  4.18m  At  watts.  ,  ,  , 

Temperature  rise  is  most  sensitively  indicated  on  a  microammeter  deflected  by  d-c 
cm-rent  generated  by  the  differential  action  of  series-connected  thermoj unctions  placed 
alternately  in  the  inlet  and  outlet  water.  The  rate  of  now  is  kept  nearly  constant  during  a 
measurement  period  by  having  a  constant-head  water  source  which  is  most  simply  a 
continuously  refilled  container  of  water  mounted  a  fixed  height  above  the  water  load. 
Calibration  of  the  overall  sensitivity  can  be  accomplished  by  dissipating  measurable  d-c 
or  low-frequency  power  in  a  resistor  which  is  immersed  in,  and  therefore  heats,  the  water. 
A  rate  of  flow  different  from  that  during  measurement  necessitates  a  proportional  correc- 
tion factor,  so  that  relative  flow  rates  must  be  checked.  Alternatively,  r-f  power  can  be 
continuously  compared  to  the  low-frequency  calibrating  power  by  using  a  balanced  bridge 
made  up  of  thermoj  unctions  in  the  water  on  each  side  of  the  calibrating  resistor  and  on 
each  side  of  the  r-f  water  load;  power  measurement  is  then  independent  of  water  flow  rate. 
Uncertainties  about  the  effects  of  spurious  heat  conduction  must  be  eliminated  by  having 
adequate  thermal  insulation  between  hot  and  cold  junctions.  Errors  from  heat  lost  by 
air  conduction,  and  thermal  conduction  down  the  line,  are  minimized  by  keeping  the 
temperature  rise  low  (order  of  1  deg  cent),  which  requires  a  high  rate  of^fiow.^  Unfor- 
tunately this  reduces  sensitivity  so  that  either  a  large  number  of  thermoj  unctions  or  a 
sensitive  microammeter  is  required  to  avoid  meter-reading  error.  For  example,  a  specific 
design  using  16  hot  and  16  cold  junctions  gave  a  60-microampere  deflection  on  a  10-ohm 
meter  for  an  input  power  of  20  watts  when  the  rate  of  flow  was  3  cc/sec. 

High-Q  matching  transformers  such  as  the  simple,  single,  quarter-wavelength  dielectric 
retainer  illustrated  in  Fig.  10,  can  match  the  water  load  to  the  transmission  line  satis- 
factorily over  only  a  narrow  frequency  band  (VSWR  within  about  1.2  over  ±3  per  cent) 
(reference  7),  and  the  internally  created  high  standing  waves  may  cause  arcing  on  high 
powers.  Low-Q,  wide-band,  matching  methods  are  usually  preferable  and  may  be  re- 
quired. One  such  method  is  to  couple  the  main  transmission  line  (coaxial  or  wave  guide) , 
by  means  of  quarter-wavelength  spaced  holes  (or  slits)  of  progressively  increasing  sizes, 
to  an  auxiliary  line  containing  a  longitudinal  dielectric  tube  through  which  the  water 
flows;  this  tube  forms  the  center  conductor  in  coaxial  line  and  is  axially  centered  in  wave 
guide.  Or  a  tapered  water  termination  can  be  simply  accomplished  in  wave  guide  by 
mounting  the  wave-guide  transmission  line  at  a  slight  angle  with  respect  to  the  horizontal. 

Accuracy.  Reflection  loss  and  other  r-f  errors  can  be  held  to  insignificant  values  by 
suitable  design.  Power-measurement  accuracy  is  governed  primarily  by  errors  in  calibrat- 
ing standards,  meter  reading,  and  thermal  loss.  These  errors  are  reduced  in  practice  to 
values  such  that  accuracies  of  0.25-0.35  db  are  commonly  attained  for  measurements  of 
power  between  20  and  100  watts. 

COMPARISON  OF  CALORIMETER  AND  BOLOMETER-ATTENUATOR  POWER 
READINGS.  A  calorimeter  and  bolometer-directional  coupler  attenuator  combination 
can  be  simultaneously  excited  by  the  same  source;  it  is  thus  possible  to  compare  readings 
directly.  Both  devices,  when  carefully  designed,  are  capable  of  such  high  accuracy  that 
cross-checks  usually  show  differences  no  greater  than  the  possible  inaccuracy  of  the 
attenuator  calibration.  If  the  calorimeter  is  the  more  reliable  it  can  be  used  to  calibrate 
the  bolometer-attenuator  combination,  which  can  then  be  used  with  equivalent  accuracy 
and  greater  convenience  to  measure  high  powers. 

34.  ATTENUATION  MEASUREMENTS 

Attenuation  is  defined  as  the  ratio  of  the  input  to  output  power  levels  in  a  network 
when  it  is  excited  by  a  matched  source  and  terminated  in  a  matched  load.  When  the 
latter  specifications  are  met  by  the  measuring  circuit,  uncertainties  in  the  measured  quan- 
tity are  avoided,  but  in  actual  use  neither  the  source  nor  the  load  may  be  exactly  matched. 
To  prevent  uncertainties  in  its  action  under  such  conditions  (see  Fig.  7),  accurately  cal- 
ibrated attenuators  are  usually  designed  to  present  nearly  matched  impedances  when 
alternate  terminals  are  terminated  with  matched  loads. 

Attenuation  is  most  accurately  measured  by  insertion  methods.  Readings  "are  obtained 
first  without  the  unknown,  then  with  the  unknown  inserted  between  a  source  and  an  r-f 
detector  (or  mixer).  To  avoid  impedance  interaction  errors  (1)  the  r-f  detector  must  be 
well  matched  (by  preceding  it  with  a  well-matched  attenuator  if  necessary) ,  (2)  the  source 
oscillator  must  be  adequately  decoupled,  and  (3)  the  source  impedance  must  be  well 
matched.  The  last  condition  can  be  assured  and  additionally  the  source  may  be  kept 
stable  during  the  measuring  period  by  monitoring  its  output  with  a  directional  coupler 
(see  Fig.  4). 


ATTENUATION  MEASUREMENTS 


11-83 


Three  methods  for  accurate  attenuation  measurement  will  be  discussed:  (1)  comparison 
against  the  calibrated  law  of  a  bolometer  power  meter,  (2)  comparison  against  a  calibrated 
i-f  attenuator  in  a  heterodyne  receiver,  and  (3)  comparison  against  a  calibrated  or  known 
r-f  attenuator. 

BOLOMETER  POWER  METER  METHOD.  The  source  is  first  terminated  by  the 
bolometer  detector,  then  by  the  attenuator  whose  output  is  terminated  by  the  bolometer 
detector.  Attenuation  is  calculated  from  the  ratio  of  the  first  to  second  power  readings 
on  the  power  meter.  Since  only  a  ratio  is  involved,  absolute  powers  need  not  be  known; 
therefore  measurement  accuracy  is  primarily  limited  by  (1)  meter-reading  accuracy,  (2) 
bolometer  detector  law  calibration,  and  (3)  temperature  (zero)  drift.  Limitation  (1)  is 
the  most  important  in  measuring  low  attenuations  (0-3  db)  for  which  accuracies  of  ±0.05 
db  are  practical.  The  accuracy  of  3-13  db  measurements  is  limited  by  (1)  and  (2)  to  about 
±0.1  db.  By  using  input  and  output  power  levels  between  10  and  Vs-Vio  mw,  attenua- 
tions of  13-20  db  are  measurable  within  ±0.1  to  0.2  db  accuracy  which  is  controlled  by 
all  three  limitations. 

HETERODYNE  RECEIVER  METHOD.  In  the  heterodyne  receiver  method  the  input 
to  the  second  detector  is  kept  constant  by  using  a  calibrated  attenuator  preceding  the 
intermediate  frequency  band-pass  amplifier.  I-f  band  widths  of  0.1  to  3  megacycles  with 
center  frequencies  of  30  or  60  megacycles  are  commonly  used.  Attenuation  is  deduced 
from  two  settings  of  the  i-f  attenuator,  one  with  and  one  without  the  unknown  between  the 
source  and  the  mixer.  Differential  frequency  stability  requirements  can  be  met  by  in- 
corporating an  automatic  frequency  control  circuit,  or,  more  simply,  they  can  be  materially 
reduced  by  using  a  frequency-modulated  mixer  oscillator.  Correct  difference  frequency 
occurs  simultaneously  with  peak  output  which  is  displayed  on  an  oscilloscope  or  a  d-c 
meter  preceded  by  a  peak  rectifier.  Sweep  methods  permit  the  alternative  use  of  video 
instead  of  i-f  amplification  and  calibrated  attenuation. 

The  minimum  signal  which  is  accurately  discernible  in  the  presence  of  noise  interference, 
and  the  maximum  signal  on  which  the  mixer  is  linear,  usually  bracket  the  maximum 
measurable  attenuation  to  about  50-70  db;  such  high-level  differences  cannot  be  accurately 
measured  unless  high-level  radiation  and  low-level  pick-up  are  eliminated  by  carefully 
shielding  all  joints  in  the  transmission  line  and  r-f  components.  For  example,  oscillator 
tubes  must  be  mounted  in  shielding  containers  into  which  power  is  supplied  through  r-f 
filters  (reference  17).  Maximum  signal  levels  for  which  the  mixer  operates  linearly  can 
be  determined  by  comparing  its  law  against  that  of  a  bolometer  power  meter  over  a  single 
or  consecutive  10-13  db  ranges.  In  the  ^.^^^^^ 

R-f  inT"    ^""-SiTa         0¥  "IR-f-OBt 


linear  region  a  power  meter,  monitoring 
and  indicating  relative  r-f  signal  levels, 
can  also  be  used  to  calibrate  or  cheek  the 
i-f  attenuator.  Attenuations  between  0 
and  60  db  are  measurable  with  an  accu- 
racy of  ±0.1-0.2  db  with  heterodyne  re- 
ceiver methods. 

R-F  ATTENUATOR  METHOD.  In 
the  calibrated  r-f  attenuator  method  all 
questions  about  linearity  are  eliminated, 
since  the  input  power  to  the  first  detector 
(or  mixer)  is  held  constant.  Attenuation 
of  the  unknown  is  deduced  from  two  set- 
tings of  the  r-f  attenuator,  one  before  and 
after  insertion  of  the  unknown. 

A  calibrated  variable  r-f  attenuator 
may  be  a  resistance-film  type  (reference 
27)  (one  for  wave  guide  is  illustrated  in 
Fig.  11),  variable  between  0  and  40  db  by 
a  precision  mechanism  which  can  be  ca- 
pable of  a  combined  setting  and  reading 
accuracy  of  ±0.1  db.  Such  attenuators 
must  be  calibrated  and  should  be  checked 
periodically  by  some  other,  more  funda- 
mental method,  which  may  introduce  an 
additional  error  of  ±0.1  db.  However, 
the  importance  of  this  lack  of  accuracy  is  often  outweighed  by  considerations  of  the  con- 
venience inherent  in  (1)  measurement  of  unknown  attenuation  by  r-f  attenuator  com- 
parison methods,  and  (2)  use  of  resistance-film  attenuators  in  them. 

The  most  stable  variable  attenuator  standards  are  of  the  wave-guide-below-cutoff  type, 


Res&fance 
film  terminations  i 

A*  Befaw-cutoff  type  for  coaxial  line: 
R-f  in  I 


B.  Below-ouioff  type  used 
with  waiteguide  lines 


Resistance  film 
C.  Resistance-f tJm  type  for  waveguide  Ooes 

FIG.  11.     Variable  Attenuators 


11-84 


ELECTRICAL  MEASUREMENTS 


and  these  are  the  only  types  for  which  incremental  attenuation  ratios  can  be  calculated 
in  advance  (references  21  and  15).  Measurement  of  unknown  attenuation  by  comparison 
against  such  an  attenuator  is  therefore  a  fundamental  method.  Unfortunately^the  min- 
imum loss,  above  which  the  attenuation  law  is  calculable  without  uncertainty,  is  around 
20-30  db  and  may  be  higher  if  input  and  output  impedances  are  corrected  by  using  masking 
attenuation  (reference  27).  Below-cutoS  attenuators  therefore  require  the  use  of  high- 
sensitivity  indicators  which  may  be  (1)  heterodyne  receivers  or  (2)  simply  low-frequency 
band-pass  amplifier  detectors  for  which  the  signal  must  be  amplitude  modulated.  The 
noise  level  may  be  higher  in  the  latter  system,  but  it  is  more  conveniently  used  since 
only  one  r-f  signal  source  is  required.  Accuracy  is  primarily  limited  by  (1)  imperfections 
in  the  attenuator  mechanism,  (2)  indicator  time  stability,  and  (3)  impedance  interaction. 
In  practice  these  factors  have  been  controlled  sufficiently  well  to  allow  ±0.2  db  accuracy 
for  0-40  db  attenuation  measurements. 


35.  FREQUENCY  MEASUREMENTS 

Two  general  methods  can  be  used  to  measure  frequency:  (1)  direct  comparison  with 
harmonics  of  a  known  frequency  source  (reference  22)  (heterodyne  frequency  meter 
method),  and  (2)  resonance  in  a  tunable  resonator  (wavemeter  method).  Except  where 
the  highest  accuracy  is  needed,  method  (2)  is  the  more  practical  since  measurements  can 
be  made  quickly  and  with  the  ramimurn  of  equipment.  A  transmission  line  propagating 
the  unknown  frequency  is  coupled  to  the  resonator  which  is  adjusted  to  resonate  at  that 
frequency;  resonance  is  indicated  by  the  response  of  a  detector  of  relative  power  level 
in  a  load  which  terminates  the  transmission  line. 

Transmission-line  resonators  can  be  constructed  to  be  self-calibrating  by  incorporating 
enough  linear  motion  so  that  the  positions  of  several  resonances  separated  by  half-wave- 
lengths can  be  measured  (lecher  wire  method).  However,  the  calibration  accuracy  is 
only  about  1  part  in  103.  The  more  accurate  practice  is  to  incorporate  just  enough  travel 
to  cover  the  desired  frequency  band  and  to  calibrate  the  resonator  by  method  (1)  above; 
in  this  manner  absolute  accuracies  of  1  part  in  10*  can  be  obtained. 

RESONATORS.  Resonators  can  be  divided  into  two  general  classes:  (1)  resonant 
transmission  lines  (coaxial  or  wave  guide) ,  and  (2)  resonant  cavities.  Resonant  transmis- 
sion lines  are  shorted  at  one  end  and 
CoBtanf  fTnger|  _Sca"le 


Antf.--backlash  Spclng 


-A..  Coaxial  resonator 


Cross-po!acJ2at)on 
suppression  loop 


Sea 


djustmg  the  line  length  to 
the  other  end  where  it  is  open  or 
short-circuited.  Resonant  cavities 
are  tuned  by  adjusting  a  reactance 
(physical  discontinuity)  in  the  elec- 
tromagnetic field.  On  transmission- 
line  resonators  the  calibrated  line 
length  changes  almost  linearly  with 
transmission-line  wavelength  (refer- 
ence 1);  one  specific  type  of  cavity 
resonator,  really  a  hybrid  combina- 
tion of  TM010  cavity  and  coaxial 
line,  can  be  constructed  to  have  a 
nearly  linear  calibration  versus  fre- 
quency over  a  band  width  as  great  as 
12  per  cent  (reference  24) . 

Table  4  illustrates  underlying  re- 
quirements to  be  met  for  accurate 
frequency  measurement  by  listing 
the  general  characteristics  of  two  of 
the  simplest  types  of  transmission- 
line  resonators  (Fig.  12) .  These  res- 
onators are  designed  for  approxi- 
mately optimum  performance  when 
diameters  are  limited  so  that  only 
the  dominant  mode  (reference  1) 

,.  (Section  7)  can  propagate.    Coaxial- 

une  resonators  are  generally  satisfactory  in  the  1500-10,000  megacycle  range*  wave- 
guide resonators  are  superior  in  the  10,000-25,000  megacycle  range  where  the  Q  of 
coaxial-line  resonators  become  too  low.  In  this  latter  range  wave-guide  resonators  using 
the  TEQ1  mode  and  special  mode  suppression  techniques  can  be  used  to  attain  higher 


Loss/  dielectric- 

masking  bearing 

contact 


lash  spun? 


S.  Te<H  moder  resonator 
FIG.  12.     Wavemeter  Resonators 


FREQUENCY  MEASUBEMENTS 


11-85 


realizable  Q's  (see  Section  7),  but  the  advantage  -will  be  slight  unless  corresponding 
improvements  can  be  introduced  to  reduce  errors  caused  by  imperfections  in  the  tuning 
mechanism. 

WAVEMETERS.  The  combination  of  a  resonator  coupled  to  a  section  of  transmission 
line  (to  which  an  unknown  frequency  source  and  a  detector  can  be  coupled)  will  be  desig- 
nated as  a  wavemeter.  Wavemeters  are  classified  according  to  the  way  the  resonator 
tuning  affects  transmission:  (1)  transmission  type,  and  (2)  suppression  type.  For  the 
first  type,  the  resonator  must  contain  two  orifices,  probes,  or  loops,  which  allow  the 
resonator  to  be  connected  in  tandem  with  the  transmission  line;  at  resonance,  power  is 
transmitted  through  the  resonator,  and  resonance  is  indicated  by  a  maximum  response  on 
the  indicator.  The  second  type  employs  a  single  orifice,  probe,  or  loop,  to  couple  the  res- 
onator's impedance  into  the  transmission  line.  The  coupled  impedance  tends  to  suppress 
the  resonant  frequency  which  is  indicated  by  a  minimum  response  on  the  detector. 

Table  4.     Typical  Wavemeter  Characteristics 


Frequency 
Band  Limits 

Theo- 
retical 
n 

Realiz- 
able 
r\ 

Possible  Errors  from 
Imperfect  Selectivity, 
Imperfect  Tuning 

Quarter 
Wave- 

Inside Dimensions: 
Diameters  and 

HJ 

Factor 

y 

Factor 

Mechanism,  Impedance 
Mismatch 

length 

Lengths 

Me 

Q 

Qcb 

Aft 
Me 

A/m 
Me 

A/z 
Me 

N 

di 
in. 

ds 
in. 

I 

in. 

Coaxial  line 

1,500 

9,200 

8,000 

0.01 

0.03 

0.01 

3 

0.65 

2.35 

5.90 

2,500 

11,500 

0.02 

0.07 

0.01 

3.54 

3,600 

7,500 

7,000 

0.04 

0.09 

0.03 

5 

0.32 

1.20 

4.14 

5,000 

8,400 

0.05 

0.2 

0.04 

2.95 

8,500 

5,800 

3,500 

0.2 

0.2 

0.1 

11 

0.16 

0.59 

3.83 

10,000 

6,200 

0.2 

0.3 

0.1 

3.24 

19,100 

4,300 

1,800 

0.8 

0.6 

0.5 

19 

0.08 

0.29 

3.10 

20,000 

4,400 

0.9 

0.7 

0.6 

2.  SO 

22,800 

3,800 

1,200 

1.5 

0.8 

1.0 

21 

0.06 

0.23 

2.72 

25,000 

3,900 

1.6 

1.0 

1.0 

2.48 

TE  1  1  mode  round 

wave  guide 

2,060 

23,000 

20,000 

0.01 

0.01 

2 

3.62 

7.3 

2,500 

25^000 

0.03 

5.6 

4,390 

19,000 

17,000 

0.02 

0.2 

0.01 

4 

1.81 

5.5 

5!  ooo 

2l!oOO 

0^6 

3.6 

9  050 

15  000 

10,000 

0.08 

0.07 

0.05 

6 

0.90 

3.6 

io!ooo 

OJ5 

2.7 

18  700 

1  1  ,000 

7,000 

0.2 

0.2 

O.I 

10 

0.45 

2.7 

20  ',000 

OM 

2.3 

72    ynn 

10  000 

5  000 

0.4 

0.3 

0.2 

12 

0.36 

2.5 

J.J  ,  /  UU 

25,000 

<K5 

2.1 

Transmission-type  wavemeters  are  not  favored  for  general  frequency  measurement, 
primarily  as  a  result  of  the  characteristic  lack  of  energy  transmission  unless  the  tuning 
is  adjusted  for  near-resonance.  Failing  to  find  a  response  on  the  detector,  the  operator  is 
faced  with  uncertainty  as  to  the  adequacy  of  the  power  output  of  the  source  or  the  sensi- 
tivity of  the  detector,  and  this  is  especially  troublesome  if  either  or  both  must  also  be 
tuned.  A  suppression  type  is  capable  of  just  as  accurate  frequency  measurements,  and 
it  is  more  convenient  to  use. 

FREQUENCY  MEASUREMENT  WITH  SUPPRESSION-TYPE  WAVEMETERS. 
The  simplest  arrangement  for  measuring  frequency  on  a  transmission  line  is  that  shown 
in  Fig.  13 A.  In  the  process  of  measuring  frequency  the  impedance  to  the  source  is  altered, 
and  this  may  change  its  frequency  unless  avoided  by  keeping  adequate  attenuation  be- 
tween the  oscillator  and  wavemeter.  Normal  operating  conditions  are  restored  by  detun- 
ing the  resonator.  In  good  designs  a  detuned  wavemeter  should  not  create  a  VSWB  in 
excess  of  1 . 1 .  This  circuit  finds  a  wide  application  where  only  a  spot  check  on  the  frequency 
of  low-power  sources  is  needed.  Figure  13B  shows  an  arrangement  for  continuously 
monitoring  frequency  with  a  negligible  disturbance  of  energy  level  and  impedance  condi- 
tions in  the  main  transmission  line.  In  some  cases  it  may  be  more  practical  to  couple  the 
auxiliary  to  the  main  transmission  line  by  means  of  a  single  orifice,  probe,  or  loop,  but  a 
directional  coupler  is  preferred  because  less  masking  attenuation  is  needed  for  the  wave- 
meter  to  operate  out  of  a  matched  impedance,  a  condition  for  which  it  is  calibrated. 


11-86 


ELECTRICAL  MEASUREMENTS 


Wavemeters  should  be  calibrated  and  operated  only  in  nearly  matched  circuits  because 
reactances  introduced  by  the  source,  or  load,  or  both,  change  the  frequency  of  minimum 
response.  Using  the  lumped-element  equivalent  circuits  in  Fig.  14  for  analysis  the 
magnitude  of  frequency  change  can  be  estimated  from:  8A/^0&  =  /(^«  +  #1),  wnere 
the  mismatched  source  and  load  introduce  impedances  having  normalized  susceptance 
components  B8  and  BI  at  the  plane  of  the  resonator  coupling.  By  using  masking  attenu- 
ators wherever  necessary,  it  is  usually  practical  to  limit  both  the  source  and  bad  impedance 
mismatch  to  1.2  VSWR;  highest  value  for  (B.  +  Bi)  =  0.4.  The  errors  in  Table  4  were 
computed  on  this  assumption  and  on  the  assumption  that  the  wavemeter  was  calibrated 
in  a  perfectly  matched  transmission  line. 

The  mechanical  system  used  for  tuning  and  indicating  the  resonant  frequency  contains 
potential'  sources  of  predominant  error.  The  error  in  interpreting  resonant  frequency 
from  a  scale  reading  can  be  estimated  from:  A/m  =  ±Mbl,  where  U  is  the  rate  of  change 
of  resonant  frequency  with  tuning  plunger  movement  in  megacycles  per  men  and  A£  is 


Oscillator      X       V |     Load 

i_j j p. 

Attenuator  Wavemeter 
A..  Spot  measuremerd: 


Source 

Probe 


A..  Probe  coupling 


Resonator 


Loacf 


Source     DirectkmaJ  coupler          Load 
J3.  Frequency  monitoring 

FIG.  13.     Frequency  Measurement  Cir- 
cuits 


Transmission 

line 
B.  Orifice  coupling 


Coupling 
orifice 


FIG.  14.     Approximately  Equivalent 
Lumped-element  Circuits 


the  maximum  possible  sum  of  (1)  backlash  between  the  driving  mechanism  and  the  scale, 
(2)  non-linearity  in  driving  mechanism  between  calibration  points,  and  (3)  uncertainty 
in  reading  the  scale.  In  Table  4,  AZ  was  assumed  as  0.0001  in.;  even  with  this  high  order 
of  mechanical  precision  the  errors  are  large.  These  errors  can  be  lowered  by  reducing  M 
(band  spreading).  In  resonant-line  resonators  this  can  be  accomplished  by  increasing  n, 
the  number  of  quarter  or  half  wavelengths.  However,  a  compromise  is  usually  made 
between  tolerable  error  and  the  frequency  range  for  which  all  resonant  settings  are  unique. 
An  alternative  method  for  band-spreading  wave-guide  resonators  is  to  operate  near  cut-off 
frequency  for  the  resonant  mode  (reference  1) ;  but  this  restricts  band  width  more  than 
the  first  method. 

Both  the  resonator's  Q  factor  and  the  tightness  with  which  it  is  coupled  to  the  trans- 
mission line  influence  the  selectivity  of  response  in  the  detector.  Using  the  circuits  in 
Fig.  14  for  analysis  it  can  be  shown  that  the  possible  error  in  tuning  the  wavemeter  result- 
ing from  inability  to  discern  the  absolute  minimum  response  on  a  power  indicator  is: 


(3) 


where  Qab  is  the  Q  of  the  total  impedance  between  terminals  a-b  (not  the  "loaded  Q") ; 
Qa&  is  always  less  than  the  theoretical  Qi  (reference  1),  and  typical  values  for  copper 
resonators,  based  upon  experimental  data  (A  =  2),  are  shown  in  the  table.  The  suppres- 
sion loss  ratio,  A,  is  the  ratio  of  the  power  in  the  load  when  the  resonator  is  completely 
detuned,  Pm,  to  the  power  in  the  load  when  the  resonator  is  tuned.  The  minimum  dis- 
cernible change  in  power  expressed  as  a  fraction  of  the  detuned  power  is  AP/Pm.  This 
equation  is  accurate  only  for  AP/Pm  below  about  10  per  cent.  Assuming  that  Pm  gives 
full-scale  deflection  and  that  a  power  change  of  0.5  per  cent  of  full  scale  can  be  discerned, 
Table  4  gives  the  possible  errors  for  some  typical  resonators. 

Differential  Accuracy.  For  well-constructed  wavemeters  the  exact  calibration  curve  is 
everywhere  smooth  and  regular.  If  terminating  reactances  remain  constant  the  accuracy 
in  measuring  small  frequency  differences  is  primarily  limited  by  the  last  two  factors  above. 


FREQUENCY  MEASUREMENTS 


11-87 


)-80OMc 
5  watts 

Microwave  frequency 
1500-25,000  Me 
—  2D  to  —40  dbm 

j         /      Directional  coupler 

L 

1  4- 

jf 

E£- 

Crystal  detector    ' 
harmonic  generator  - 

Directional 
coupler    - 

"  p^.  ¥M    Crystal 
-o                     mixer 

\    § 
•A    c 

*   ~Z                                           Crystal 
-<           ,      /"""N                detector 
)  Attenuator  ^    J  Attenuator 

, 

m, 

•  ""[    .                                       &Z&&7                      -^S^^f 

""LJ 

1 

Klystron  local      * 
oscillator 

"*-•  —  Repeller 

Wavemeter 

under  test 

| 

Sawtooth 
sweep 

—3 

^    Oscillos- 

COpe 

2-5QJ<c 
amplifier 

« 

hign^gam 

Each  one  of  these  errors  can  enter  twice.  According  to  Table  4,  differential  measurements 
can  be  made  with  an  accuracy  no  worse  than  about  1  part  in  104.  Accuracies  of  5  in  10s 
are  commonly  realized  in  practice. 

"Wavemeter  Calibration.  Three  somewhat  different  methods  can  be  used  to  calibrate 
the  wavemeter  against  low-frequency  standards.  In  the  first  method  the  wavemeter  is 
adjusted  to  resonate  at  a  c-w  frequency  supplied  by  a  local  microwave  oscillator;  this 
frequency  is  then  measured  with  a  precision  heterodyne  frequency  meter  (accurate  to  1 
part  in  106)  such  as  shown  in  reference  22. 

In  the  second  method,  microwave  frequencies  standards  are  harmonically  generated 
from  low-frequency  standards;  a  calibration  is  then  made  by  adjusting  the  frequency*meter 
to  resonate  on  these  standards  (reference  27).    The  response  must  be  detected  in  a  high- 
gain  low-noise  detector  such  as  a  dou- 
ble-detection receiver,  since  the  har- 
monics power  output  is  very  low. 

The  third  method  is  illustrated  in 
Fig.  15,  which  combines  advantageous 
features  of  both  the  above  methods. 
The  microwave  oscillator  is  frequency 
modulated  by  a  saw-tooth  wave  on  its 
repeller  (references  9  and  10) .  Its  cen- 
ter frequency  is  adjusted  until  a  low- 
amplitude  marker  oscillation  appears 
superimposed  on  the  rectified  envelope 
of  the  oscillator  output;  this  pip  ap- 
pears during  the  time  interval  for 
which  the  frequency  zero-beats  (2-50 
kc)  with  a  standard  frequency  gener- 
ated by  the  crystal  rectifier  multi- 
plier. The  frequency  meter  is  tuned 
until  its  null  brackets  the  marker  pip. 
Frequency-time  stability  of  the  f-m 
oscillator  is  unimportant,  and  the  low- 
level  output  from  the  harmonic  gener- 
ator is  used  only  for  frequency  mark- 
ing. Additionally,  a  known  c-w  cali- 
brating frequency  can  be  obtained  by 
gradually  reducing  the  sweep  to  zero 
while  keeping  the  marker  in  the  center  of  the  mode;  a  beat  note  in  the  head  phones  can 
be  tuned  in  and  retained  by  trimming  adjustments  on  the  frequency. 

Conventional  methods  employing  a  master  crystal-controlled  oscillator  and  vacuum- 
tube  multipliers  can  be  used  for  generating  the  standard  frequencies  up  to  800  megacycles 
(see  p.  7-92).  These  frequencies  are  then  multiplied  20  to  60  times  in  a  silicon  crystal 
rectifier  mounted  in  the  microwave  transmission  line.  The  harmonic  number  can  be  easily 
identified  by  a  roughly  calibrated  or  self-calibrating  wavemeter.  High  burn-out  crystals 
(reference  6)  are  recommended  since  they  handle  more  input  power.  The  microwave 
signal  power  so  produced  is  of  the  order  of  -20  to  -50  dbm.  A  severe  requirement  is 
therefore  placed  upon  the  noise  figure  of  the  high-gain  detection  system  used  to  amplify 
the  response  to  a  usable  amplitude. 

Absolute  Accuracy.  By  the  above  methods  standard  frequencies  can  be  known  to  an 
accuracy  of  1  part  in  106.  Adding  to  this  the  accuracy  with  which  these  standard  frequen- 
cies can  be  transferred  to  the  wavemeter  calibration  curve  (limited  primarily  by  the  first 
two  errors  listed  in  Table  4)  results  in  a  typical  absolute  calibration  accuracy  of  about  5 
parts  in  105.  Then,  assuming  that  corrections  for  temperature  and  humidity  can  be  made 
with  negligible  error,  adding  aU  three  errors  in  Table  4  to  the  calibration  accuracy,  a  typical 
wavemeter  can  be  used  to  measure  frequency  with  an  absolute  accuracy  of  1-2  parts  in  10  . 
Correction  for  Temperature  and  Humidity.  In  order  to  interpret  the  resonant  frequency 
from  a  scale  reading  and  the  calibration  chart  accurately,  corrections  must  be  made  to 
allow  for  the  effects  of  any  change  from  the  reference  temperature  and  humidity  conditions 
under  which  the  resonator  was  calibrated.  Two  effects  operate  independently,  and  the 
corrections  for  them  can  be  computed  separately,  then  added  algebraically:  (1)  tempera- 
ture changes  induce  thermal  expansion  or  contraction  of  the  resonator's  internal  dimen- 
sions; (2)  combined  temperature  and  relative  humidity  changes  vary  the  dielectric  constant 
of  the  air  inside  unsealed  resonators  (reference  28). 

The  correction  for  thermal  expansion  of  homogeneous  resonators  can  be  computed  from: 
A/  =  —  C/o  A£,  where  A/ is  the  frequency  correction  to  be  added  to  the  resonant  frequency 


Head  phones 

Harmonic       j  Spread 

marker         j  scale 

Wavemef  er  tuned 

FIG.  15.     Wavemeter  Calibration  Circuit 


11-88 


ELECTRICAL  MEASUREMENTS 


/o  indicated  by  the  calibration  chart,  Ar  is  the  change  in  temperature  from  that  for  which 
the  calibration  chart  was  prepared,  and  C  is  the  coefficient  of  linear  expansion  for  the 
resonator  material.  The  correction  is  small  for  resonators  made  of  Invar;  A///O  is  about 
d=5.5  X  10  ~7  for  ±1  deg  fahr  temperature  change.  For  the  same  temperature  change 
A///O  is  about  ±6.6  X  10~6  for  steel  resonators,  and  ±1  X  10"5  for  brass  resonators. 

Corrections  for  the  effect  of  dielectric  constant  changes  are  given  in  Table  5,  which 
applies  for  unsealed  resonators  operated  at  sea-level  atmosphere  (reference  27).  The 
corrections  are  to  be  algebraically  added  to  the  indicated  resonant  frequency  (assuming 
that  the  calibration  was  made,  or  normalized,  for  25  deg  cent  and  60  per  cent  relative 
humidity  conditions).  It  will  be  observed  that  for  ±10  per  cent  change  in  relative 
humidity  the  correction  is  dbl.3  X  10 "^  at  25  deg  cent. 

Table  5.     Resonant-frequency  Correction  for  Humidity  Changes  in  Unsealed  Resonators 


Temper- 
ature, 
deg  cent 

Relative  Humidity,  per  cent 

Temper- 
ature, 
deg  fahr 

20 

30 

40 

50 

60 

70 

80 

90 

100 

0 

+0.0050 

+0.0045 

+0.0042 

+0.0050 

+0.0038 

+0.0035 

+0.0032 

+0.0028 

+0.0025 

32 

5 

+0.0052 

+0.0048 

+0.0042 

+0.0040 

+0.0035 

+0.0032 

+0.0027 

+0.0022 

+0.0018 

41 

10 

+0.0055 

+0.0049 

+0.0042 

+0.0038 

+0.0031 

+0.0025 

+0.0020 

+0.0013 

+0.0008 

50 

15 

+0.0055 

+0.0048 

+0.0041 

+0.0032 

+0.0024 

+0.0015 

+0.0010 

+0.0001 

-0.0006 

59 

20 

+0.0055 

+0.0045 

+0.0035 

+0.0025 

+0.0015 

+0.0005 

-0.0006 

-0.0015 

-0.0025 

68 

25 

+0.0055 

+0.0041 

+0.0028 

+0.0013 

+0.0000 

-0.0013 

-0.0025 

-0.0040 

-0.0055 

77 

30 

+0.0050 

+0.0032 

+0,0025 

-0.0003 

-0.0018 

-0.0038 

-0.0052 

-0.0071 

-0.0080 

86 

35 

+0.0045 

+0.0023 

+0.0002 

-0.0022 

-0.0045 

-0.0067 

-0.0089 

-0.0100 

-0.0133 

95 

40 

+0.0030 

+0.0010 

-0.0018 

-0.0047 

-0.0075 

-0.0105 

-0.0131 

-0.0158 

-0.0188 

104 

45 

+0.0027 

-0.0007 

-0,0043 

-0.0078 

-0.0115 

-0.0150 

-0.0186 

-0.0221 

-0.0256 

113 

50 

+0.0014 

-0.0031 

-0.0075 

-0.0119 

-0.0163 

-0.0209 

-0.0252 

-0.0295 

-0.0340 

112 

Frequency  correction,  per  cent  to  be  added  to  indicated  reading,  for  resonators  calibrated  at  25  deg  cent  and  60  per 
cent  relative  humidity. 

Frequency  Tuning.  Suppression  loss  ratios  lower  than  2  increase  the  difficulty  in 
finding  the  region  of  resonance  when  the  response  is  indicated  on  a  d-c  meter;  since  meters 
are  inherently  sluggish  the  narrow  resonant  region  can  be  easily  missed.  For  example, 
the  resonant  region  in  terms  of  band  width  between  the  two  frequencies  for  which  the  power 
change  is  half  the  total  null  change  is  only  A/Q0&  =  /VZ".  On  wide-range  wavemeters  it 
is  often  desirable  to  keep  this  resonant  region  relatively  wide  by  purposely  designing  for 
only  a  moderate  value  of  Qab- 

High-suppression  moderate-Q  requirements  are  relatively  unimportant  when  a-m 
(including  pulsed)  wave  envelopes  are  displayed  on  an  oscilloscope  screen.  Since  oscillo- 
scopes respond  to  rapid  level  changes  a  reaction  can  usually  be  discerned,  even  though  the 
resonant  region  be  tuned  through  rapidly.  In  some  cases  suppression-loss  ratios  as  low 
as  1.1  to  1.2  may  be  desirable  in  order  to  avoid  excessive  distortion  of  the  oscilloscopic 
pattern  in  the  region  of  resonance. 

The  oscilloscope  pattern  in  Fig.  16  illustrates  the  superposition  of  a  wavemeter  null 
("pip")  at  two  frequencies  on  the  envelope  of  the  output  of  a  klystron  oscillator  which  is 
being  frequency  and  amplitude  modulated  by  a  sawtooth  voltage 
on  its  repeller  (reference  9).    The  "pip"  can  so  easily  be  located 
on  this  type  of  wave  that  auxiliary  facilities  comprising  an  adjust- 
\  J  able  sweep  voltage  and  a  crystal  detector  with  oscilloscope  indi- 

X /  cator  provide  a  convenient  method  for  quickly  tuning  klystron 

FIG.   16.     Wavemeter   sources  to  required  c-w  frequencies  (as  set  on  the  wavemeter). 
"Pips"  on  Klystron  Mode    The  initial  sweep  voltage  is  large  enough  to  sweep  through  an  oscil- 
lation mode  completely,  regardless  of  the  initial  settings  of  the 

oscillator  tuning  controls;  tuning  is  adjusted  until  the  wavemeter  pip  appears  at  the  top 
of  the  mode.  The  sweep  voltage  is  then  gradually  reduced  to  zero  and  the  oscillator 
tuning  controls  are  simultaneously  adjusted  to  keep  the  "pip"  centered  on  the  mode. 


BIBLIOGRAPHY 

1.  Schelkunoff,  S.  A.,  Electromagnetic  Waves,  D.  Van  Nostrand  Co.,  New  York,  1943 
o    iiven    *  '  Communication  Engineering,  McGraw-Hill  Book  Co    New  York   1937 

A    Q  ° StPr?^*  V*   T  *  Technique*  of  Microwave  Measurements,  McGraw-Hill,  New' York   1947 
4.  Smith,  P.  H.,  An  Improved  Transmission  Line  Calculator,  Electronics  January  1944 

n"*"   Publication  23-80,  Sperry  Gyroscope  Co.,  1944. 
Uectronic$r  July  1946,  p.  112. 


OSCILLATORS  FOR  SIGNAL  GENERATOR  USE         11-89 


?'  ^^yton,  R.J.,  et  al.,  Radio  Measurements  in  the  Decimeter  and  Centimeter  Wavebands,  /.  I.E.E.J 

8.  Becker,  J.  A     et  al.,  Properties  and  Uses  of  Thermistors— Thermally  Sensitive  Resistors,  BeU 

bys.  Tech.  J.,  January  1947. 

9.  Pierce,  J.  R    Reflex  Oscillators,  Proc  I.R.E.,  Vol.  33,  112  (February  1945). 

10.  Genzton,  E.  L    and  Harrison,  A.  E.,  Reflex-klystron  Oscillators,  Proc.  I.R.E.,  VoL  34  (March  1946). 

11.  iiske,  J.  B.,  et  al.,  The  Magnetron  as  a  Generator  of  Centimeter  Waves,  Bell  Sys.  Tech.  J.,  Vol.  25 

(.April  1946) . 

12.  HasseU, FM  and  Jenks,  F.,  Electroforming  Microwave  Components,  Electronic^  March  1946. 

}*  ££      i     J1  ^ral^rProbe  Error  m  Standing-wave  Detectors,  Proc.  I.R.E.,  Vol.  34,  33P  (January  1946). 

14.  Mumford,  W.  W.,  Directional  Couplers,  Proc.  I.R.E.,  Vol.  35,  160  (February  1947). 

J5*  S^?n'  E-  IV  «  al"  Microwave  Radar  Testing,  Trans.  AJ.E.E.,  Vol.  65,  274  (May  1946). 

16.  Whinnery,  J.  R.T  et  al.,  Coaxial  Line  Discontinuities,  Proc.  I.R.E.,  Vol.  32,  695  (1944). 

17.  Ragan,  G.  L.,  Microwave  Transmission  Circuits,  McGraw-Hill,  New  York,  194S. 

18.  Tyrrell,  W.  A.,  Hybrid  Circuits  for  Microwaves,  Proc.  LR^E.,  Vol.  35  (November  1947). 

19.  Type  821  Barretter,  Technical  Data  Bulletin,  Sperry  Gyroscope  Co. 

20.  Guillemin,  E.  A.,  Communication  Networks,  John  Wiley  &  Sons,  New  York,  1949. 

21.  Lindner,  E.  G.,  Attenuation  of  Electromagnetic  Fields  in  Pipes  Smaller  Than  Critical  Size,  Proc. 

I.R.E.,  Vol.  30,  554  (December  1942). 

22.  Essen,  L.,  and  Gordon-Smith,  A.  G.,  The  Measurement  of  Frequencies  in  the  Range  100  Mc/s  to 

10,000  Mc/s,  J.  I.E.E.,  Vol.  92,  Part  III,  291  (December  1945). 

23.  Essen,  L.,  The  Design,  Calibration,  and  Performance  of  Resonance  Wavemeters  for  Frequencies 

between  1000  and  25,000  Mc/s.,  J.  I.E.E.,  Vol.  93,  Part  IIIA.,  No.  9,  1946. 

24.  Essen,   L.,   Cavity-resonator  Wavemeter — Simple  Types  of  Wide  Frequency  Range,    Wireless 

Engineer,  Vol.  23,  126  (May  1946). 

25.  Swedin,  M.,  Directive  Couplers  in  Wave  Guides,  J.  I.E.E.,  Vol.  93,  Part  IIIA,  No.  4,  1946. 

26.  Rosen,  S.,  and  Bangert,  J.  T.,  A  Consideration  of  Directivity  in  Wave  Guide  Directional  Couplers, 

Proc.  I.R.E.,  Vol.  37  (April  1949). 

27.  Gaffney,  F.  J.,  Microwave  Measurements  and  Test  Equipments,  Proc.  I.R.E.,  Vol.  34,  775  (October 

28.  Englund,  C.  R.,  et  al.,  Further  Results  of  a  Study  of  Ultra-short-wave  Transmission  Phenomena, 

Bell  Sys.  Tech.  J.,  Vol.  19,  369  (July  1935). 


SIGNAL   GENERATORS  AND  POWER   MEASUREMENT 

By  F.  J.  GafEney 

A  signal  generator  is  a  source  of  alternating  voltage,  calibrated  in  frequency  and  voltage 
(or  power  output  into  a  specified  load),  and  with  good  modulation  characteristics,  and 
carefully  shielded.  Units  are  commercially  available  which  collectively  cover  frequency 
ranges  from  audio  to  microwave  frequencies.  Designs  are,  in  general,  made  as  broad  band 
as  the  limitations  of  oscillator-tube  and  power-output  calibration  devices  (including  the 
output  attenuator)  will  allow.  Problems  encountered  in  the  design  of  signal  generators 
are:  (1)  the  design  of  stable  oscillator  circuits,  (2)  the  design  of  systems  of  modulation,  (3) 
output  voltage  or  power  standardization,  (4)  attenuator  design,  (5)  shielding. 

36.  OSCILLATORS  FOR  SIGNAL  GENERATOR  USE 

Signal  generator  oscillators  are  designed  for  maximum  frequency  stability  with  tempera- 
ture and  line  voltage.  Wherever  possible  it  is  desirable  to  load  the  oscillator  lightly.  For 
this  reason  it  is  advantageous  at  some  frequencies  to  employ  buffer  amplifiers  which  feed 
the  output  circuits  rather  than  to  feed  these  circuits  directly  from  the  signal  generator 
oscillator  itself. 

Tubes  and  Circuits.  Any  oscillator  consists  essentially  of  a  tuned  amplifier  with  suffi- 
cient positive  feedback  to  supply  the  grid  losses.  The  frequency  stability  depends  on  the 
frequency  shift  necessary  to  restore  proper 
phase  in  the  grid  circuit  when  the  tube  char-  Ganged  CQndens 

acteristics  change  as  the  result  of  line  voltage,      -^ 

thermal  effects,  etc.    This  frequency  shift  is  a         \  .^wvvlf         ?        »  Output 

function  of  the  Q  of  the  tuned  circuit  em-  v          ^^ 

ployed,  of  the  feedback  system  used,  and  of 
the  method  of  coupling  to  the  tube. 

AUDIO-FREQUENCY      OSCILLATORS. 
For  the  range  from  a  few  cycles  to  20  kc  or  ^"T~*  ^Biasing  lamp 

more,  various  forms  of  rC  oscillators  have  1 

proved  advantageous.    This  type  of  oscillator  -i 

is  essentially  an  untuned  amplifier  with  an  p^  ^    wien  Bridge  Oscillator 

rC  filter  providing  the  tuned  feedback.    The  _ 

filter  may  be  of  the  twin  T  or  Wien  bridge  type.    A  diagram  of  the  latter  is  shown  in  Fig.  1. 

"  >  sharpness  of  the  filter,  which  in  turn  depends  on 


11-90 


ELECTRICAL  MEASUREMENTS 


/,-£ 

R-f 

iUier 

Audio 
ajnplifier 

the  tracking  of  the  tuning  condensers.  Means  are  usually  incorporated  for  providing  an 
automatic  bias  on  the  oscillator  tube  to  insure  Glass  A  operation  as  the  oscillator  output 
changes  over  the  frequency  band.  This  can  take  the  form  of  a  non-linear  resistance  such 
as  a  tungsten  lamp  in  the  cathode  circuit  of  the  oscillator.  This  type  of  oscillator  if  properly 
designed  can  be  made  extremely  stable  and  pure  in  waveform.  One  disadvantage  is  the 
inability  of  a  single  tuning  condenser  to  cover  a  large  frequency  range.  For  this  reason  a 
step  switch  is  usually  provided  which  switches  the  resistances  in  the  circuit  to  provide 
multiple  ranges. 

For  wider-range  low-frequency  oscillators  (from  a  few  cycles  to  several  megacycles  per 
second)  beat-frequency  oscillators  may  be  used.  A  block  diagram  of  such  an  oscillator  is 
shown  in  Fig.  2.  It  consists  essentially  of  two  r-f  oscillators,  one  fixed  and  one  variable, 


Output  to 
"  attenuator 


FIG.  2.     Block  Diagram  of  Beat  Frequency  Oscillator 

which  feed  a  mixer  tube.  The  difference  frequency  between  the  r-f  oscillators  is  then  fed 
through  an  r-f  niter  to  an  audio  amplifier.  This  type  of  circuit  has  the  advantage  that  a 
very  large  frequency  range  can  be  covered  with  small  changes  in  the  frequency  of  the  vari- 
able r-f  oscillator.  Its  main  disadvantage  is  that  the  audio  frequency  is  the  difference  be- 
tween two  radio  frequencies  so  that  a  small  percentage  variation  in  one  of  these  frequencies 
will  produce  a  relatively  large  variation  in  the  output  frequency.  However,  both  r-f  oscil- 
lators can  be  designed  so  as  to  be  very  similar  in  construction,  and  under  these  conditions 
they  have  a  tendency  to  drift  together  so  as  to  minimize  the  drift  in  the  audio  frequency.  A 
serious  problem  which  exists  in  the  design  of  this  type  of  oscillator  is  concerned  with  spuri- 
ous outputs  produced  by  the  beating  of  harmonics  of  the  r-f  oscillators.  These  effects  can 
be  minimized  by  utilizing  pure  r-f  waveforms,  by  the  use  of  suitable  filters  between  r-f 
and  mixer  stages,  and  by  careful  mixer  design  for  large  signal  mixing. 

RADIO-FREQUENCY  OSCILLATORS.  For  the  range  of  frequencies  up  to  about  100 
megacycles  per  second,  conventional  lumped  constant  circuits  may  be  used.  Several  types 
of  circuits  such  as  those  described  in  Section  7  may  be  employed.  In  order  to  obtain  good 
frequency  stability,  either  impedance-stabilized  oscillators  or  oscillators  of  the  electron- 
coupled  type  should  be  used.  With  the  latter  type  of  oscillator,  frequency  variation  de- 
pends on  the  ratio  of  the  screen-grid  voltage  to  the  plate  voltage,  and  for  some  value  of 
this  ratio  the  frequency  variation  is  extremely  small  with  variations  in  plate  voltage. 
The  electron-coupled  oscillator,  though  good  from  the  standpoint  of  stability,  produces  a 
poor  waveform  which  contains  many  harmonies.  Care  must  therefore  be  taken  in  the  use 
of  such  an  oscillator  to  insure  that  the  harmonic  content  does  not  affect  the  power-measur- 
ing circuit  or  the  receiver  being  tested.  The  frequency  stability  of  all  types  of  oscillators 
is  improved  by  regulation  of  the  B  voltage.  Some  frequency  instability  can  be  produced 
by  variation  in  cathode  heater  voltage,  but  this  is  seldom  compensated  for  in  practical 
design.  Frequency  stability  with  variation  in  ambient  temperature  is  accomplished 
through  careful  design  of  the  tuned  circuit  of  the  oscillator.  Coil  forms  should  be  wound 

on  a  material  having  a  very  low  coefficient  of  expansion, 
such  as  Vicor.  Special  attention  must  be  paid  to  the  de- 
sign of  the  variable  condenser  to  minimize  dimensional 
changes  with  temperature. 

At  frequencies  above  100  megacycles  per  second  con- 
siderable difficulty  is  encountered  in  the  use  of  conven- 
tional lumped  circuits.  The  losses  in  such  circuits  be- 
come excessive  at  these  frequencies,  and  the  tuning  range 
becomes  small,  since,  as  the  inductance  is  decreased  to 
obtain  higher  frequencies,  the  tube  and  wiring  capaci- 
tances become  a  larger  proportion  of  the  total  allowable 
capacitance.  One  solution  to  this  problem  has  been  in 
the  use  of  circuits  of  the  butterfly  type  such  as  that  shown 
in  Fig.  3.  Here  the  inductance  and  capacitance  are  varied  simultaneously.  This  has 
the  effect  of  increasing  the  tuning  range  and  simultaneously  maintaining  the  L/C  ratio. 
Such  circuits  have  been  built  for  frequencies  up  to  2000  megacycles  per  second.  At 
frequencies  above  1000  megacycles  per  second,  however,  the  losses  are  too  great  for 
practical  oscillator  applications. 


FIG.  3.     Butterfly  Tuning  Circuit — 

400  to  1200  megacycles  per  second 

(Courtesy  of  General  Radio  Co.) 


OSCILLATORS  FOR  SIGNAL  GENERATOR  USE         11-91 

in   heo?™?11 10  ^  Pr°bl6m  °f  obtaitting  toned  circuits  at  high  frequencies  consists 
Here  Sati™ Tff        T*  -°f  tr*nsmission  ^  «  in  the  use  of  cavity-type  resonators. 
'  t«  flow         T   ^  ehmmated  and  the  distributed  nature  of  the  circuit  allows  cur- 
nne  lenTh  fT  f^'  ?""?  Permitt^  1™  copper  losses.    For  a  lossless  trans- 
hne length  I,  short-circuited  at  the  far  end,  the  input  impedance  *,-„  is  given  by: 


.  ATTL 

•  ZQ  tan  — — 

A 


(1) 


*°  .77cllaracteristic  impedance  of  the  Hne  and  X  =  wavelength 

«nHSf  n~f  /  4'  \=  u°°.'  ACtUally'  Owing  to  losses'  transmission  lines  dissipate  energy 
and  have  a  Q  factor  which  is  defined  in  terms  of  band  width  as  in  a  lumped  constant  circuit. 
.transmission  line  Q  is  approximately  given  by: 


where/o  =  frequency  of  resonance  (cycles  per  second),  ^  =  characteristic  impedance  = 
V  L/C  (ohms),  r  =  resistance  per  unit  length  (ohms/meter),  and  c  =  velocity  of  light 
(meters  per  second)  . 

The  input  impedance  of  a  quarter  wavelength  short-circuited  line  is  then  approximately 
given  by: 


In  the  equation  for  Q,  both  SQ  and  r  vary  with  the  dimensions  of  the  hne.  For  maximum  Q 
there  exists  an  optimum  diameter  ratio  for  coaxial  lines  (for  conductors  of  the  same  resis- 
tivity) amd  an  optimum  ratio  of  spacing  to  the  diameter  of  the  conductors  for  parallel 
wire  lines!  Letting  this  ratio  be  b/a  for  the  two  cases,  one  value  of  b/a,  gives  maximum  Q 
and  a  second  value  gives  maximum  input  impedance.  The  values,  together  with  the 
corresponding  characteristic  impedances  are  given  in  the  table. 


COAXIAL  LINES 


b/a 

Max.  Q.. ., 3.6 

Max.  z 9.2 


zo 

76.8 
133,1 


5/o 
4.0 
8.0 


WISE  LINES 

20 

851. 
1934. 


With  b/a  constant,  both  Q  and  z  increase  linearly  with  b.  Such  a  line  can  then  be  used 
as  an  antiresonant  circuit,  with  the  resonance  frequency  determined  by  the  line  length. 
Actually  the  shapes  of  the  curves  of  reactance  and  resistance  as  a  function  of  frequency 
are  not  identical  to  those  of  lumped  constant  circuits  but  are  quite  similar  near  the  resonant 
frequency.  Parallel  wire  transmission  lines,  because  of  then*  simplicity  of  construction, 
are  sometimes  used  in  experimental  oscillators,  but  coaxial  transmission  lines  are  more 
commonly  used  in  signal  generator  applications  because  of  their  lower  losses  (the  radiation 
•  loss  being  zero)  and  self-shielding  construction.  The  only  difficulty  with  such  resonant 
lines  is  in  the  physical  lengths  required  and  in  the  mechanical  difficulties  with  sliding 
contacts.  For  a  frequency  of  100  megacycles  per  second,  for  instance,  a  quarter-wavelength 
transmission  Hne  would  have  a  length  of  75  cm.  The  Hne  may,  however,  be  artificially 
shortened  by  the  use  of  a  fixed  condenser  across  its  input  terminals. 

Vacuum  tubes  designed  for  low-frequency  appHcations  fail  at  higher  frequencies  because 
of  losses,  Hmitations  due  to  input  and 

output  capacitances,  and  transit  time  [< — — 1« H  Alternate  method  PutJ 

effects.     To  minimize  these  defects,  $$  (  ^f  output  coupling       f 

special  tubes  have  been  designed  for 
the  higher-frequency  ranges.  By  mak- 
ing the  tube  elements  very  small,  spac- 
ings  between  elements  can  also  be 
made  small  without  the  introduction 
of  excessive  interelectrode  capaci- 
tances. Lead  inductances  may  be 
minimized  by  bringing  out  more  than 
one  lead  from  each  electrode  as  in  some 
types  of  high-frequency  acorn  tubes. 
A  better  scheme  is  that  exemplified  by  tubes  of  the  2C40  type  which  utilize  disk  seal 
construction  to  reduce  lead  inductances  to  a  minimum.  This  type  of  oscillator  tube  lends 
itself  particularly  well  to  incorporation  into  a  coaxial-Hne  oscillator  circuit.  One  such 
design  showing  a  tuned  plate-tuned  grid  oscillator  is  shown  in  Fig.  4.  Owing  to  the 
difference  in  end  effects  and  to  the  capacitive  loading  contributed  by  the  tube  elements, 


FIG.    4. 


Input 


Typical    Double-concentric-eavity    Circuit 
(Courtesy  of  General  Electric  Co.) 


11-92 


ELECTRICAL  MEASUREMENTS 


it  is  usually  necessary  to  move  the  tuning  plungers  at  a  different  rate.   This  causes  some 

difficulty  with  mechanical  tracking  of  the  two  cavities. 

An  alternative  design  known  as  a  reentrant  oscillator  is  shown  in  Fig.  5.    This  type  of 

oscillator  may  be  controlled  by  means  of  a  single  tuning  plunger  but  will  operate  satis- 
factorily only  over  a  relatively  narrow  fre- 
quency band.  Oscillators  using  this  type  of 
circuit  with  a  2C40  triode  tube  have  been  made 
for  frequencies  as  high  as  3000  megacycles  per 
second.  Experimental  tubes  have  been  made 
to  oscillate  in  this  type  of  circuit  at  even  higher 
frequencies.  Because  of  the  close  grid  cathode 

FIG.  5.    Method  for  Tuning  Re-entrant  Oscil-  spacing  a  considerable  frequency  variation  may 

lator  Circuit  (Courtesy  of  General  Electric  Co.)   be  produced  by  variations  of  the  tube  heater 

voltage. 
At  frequencies  above  2000  megacycles  per  second,  velocity  variation  tubes  become  useful. 

In  this  type  of  oscillator  tube  transit  time  is  utilized  to  provide  bunching  of  the  electrons 

in  a  drift  space.    The  efficiency  of  this  type  of  oscillator  is  low  and  the  frequency  stability 

is  relatively  poor.    No  upper  frequency  limit  exists  for  oscillators  of  this  type  except  that 

imposed  by  problems  of  physical  construction.    This  limitation  occurs  somewhere  in  the 

region  of  60,000  megacycles  per  second.  . 

Cavity  resonators  which  may  either  be  fj) 

external  to  the  tube  or  integral  with  it  '  ' 

are  utilized  with  this  type  of  oscillator. 

The  type  of  cavity  resonator  used  is  one 

which  develops  maximum  voltage  across 

the  bunching  grids  in  the  tube. 

The  type  of  velocity  variation  tube 

known  as  a  reflex  oscillator  is  most  con- 
venient for  signal  generator  applications. 

An  outline  diagram  of  this  type  is  shown 

in  Fig.  6.     In  this  tube  the  electron 

stream  is   velocity  modulated  by  the 

bunching    grids.      The    electrons   then 

drift  in  a  space  between  the  bunching 

grids  and  a  negatively  charged  reflector 

which  turns  them  around  and  causes 


Output  coupling' 
loop 


,,Tlmhig  plunger 
.Reflector 


. ^Drlft  space 


'Resonator  ana's 
•Eleclfoj5  gun 


•Glass  envelope 


FIB.  6.    Reflex  Type  Velocity  Variation  Oscillator 


bunches  to  arrive  again  at  the  bunching  grids  in  such  phase  as  to  deliver  energy  to  the 
cavity  resonator  of  which  the  bunching  grids  form  a  part.  The  resonant  frequency  of  such 
an  oscillator  may  be  changed  by  varying  the  cavity  dimensions.  Alternatively,  the  fre- 
quency may  be  changed  by  changing  the  capacitance  between  the  buncher  grids.  The 
latter  method  requires  that  the  tube  be  built  with  a  flexible  diaphragm  since  the  grids  are  of 


Fro.  7.     TM-mode  Cavity  for  Reflex  Velocity  Variation  Oscillator 


course  in  an  evacuated  space.  The  realizable  tuning  range  with  this  type  of  tuning  is  only 
about  20  per  cent.  This  design  does,  however,  possess  the  advantage  that  the  tube  and 
the  circuit  are  self-contained  and  troubles  with  sliding  contacts  are  eliminated. 

By  bringing  the  bunching  grids  out  through  disk  seals  in  a  glass  envelope   an  external 
eavity  may  be  used  with  this  type  of  tube.    Cavities  of  both  coaxial  mode  and  the  TM 


MODULATION  OF  SIGNAL  GENERATORS 


11-93 


mode  have  been  employed  successfully.  A  cavity  of  the  coaxial-mode  type  is  shown  in 
±"ig.  t>;  .tig.  7  snows  an  outline  drawing  of  the  TM-mode  cavity.  With  the  coaxial-type 
cavity  frequency  ranges  of  better  than  2  to  1  may  be  achieved,  Care  is  again  necessary 
in  the  construction  of  tuning  plungers  so  as  to  make  good  contact  over  the  tuning  range. 
In  order  to  minimize  rubbing  contact,  it  is  possible  to  design  choke-type  contacts  which 
are  themselves  resonant  sections  of  transmission  lines. 


ibilizing  /-Audio  bypass 

/.   condenser 


, ^Stabiliz 

/         //Condenser  ,'j_ 

•Xr — r^W-C* 

R-f  choke  f 


37.  MODULATION  OF  SIGNAL  GENERATORS 

In  order  to  test  the  detection  characteristics  of  receivers  and  to  supply  a  signal  that  can 
be  amplified  by  audio  amplifier  methods,  signal  generators  are  usually  equipped  with  means 
for  modulating  the  r-f  voltage  output.  Several  types  of  modulation  are  used,  depending 
on  the  types  of  equipment  with  which  the  signal  generator  is  most  likely  to  be  employed. 
Signal  generators  for  use  in  the  a-m  broadcast  band,  for  instance,  are  provided  with  ampli- 
tude modulation,  and  those  for  use  hi  testing  f-m  receivers  are  frequency  modulated.  It 
is  generally  desirable  to  limit  the  modulation  to  one  type  and  to  provide  means  of  reducing 
unwanted  types  of  modulation. 

AMPLITUDE  MODULATION.  Both  the  impedance-stabilized  oscillator  and  the 
electron-coupled  oscillator  are  susceptible  to  plate  modulation  with  little  accompanying 
frequency  modulation.  In  the  impedance-stabilized  oscillator,  the  frequency  is  inde- 
pendent of  applied  plate 
voltage  over  a  wide  range 
if  the  proper  value  of  sta- 
bilizing impedance  is 
used.  Since  signal  gener- 
ator oscillators  are  tuned 
over  wide  frequency 
ranges,  however,  it  is  nec- 
essary to  track  the  stabi- 
lizing impedance  with  the 
main  frequency  control  if 
satisfactory  performance 
is  to  be  obtained.  In  the  Hartley  oscillator,  the  stabilizing  impedance  is  a  condenser  for 
both  the  grid  and  plate  stabilization  types,  and  this  condenser  must  be  kept  proportional 
to  the  total  value  of  tuning  condenser  as  the  latter  is  varied.  Figure  8  is  a  diagram  showing 
this  type  of  oscillator  with  plate  modulation. 

The  audio  power  output  from  the  modulator  tube  must  be  about  3  times  the  r-f  power 
from  the  oscillator  if  100  per  cent  modulation  is  to  be  obtained.  This  can  be  accomplished 
by  means  of  a  voltage  dropping  resistor  between  the  plates  of  the  r-f  and  modulator  tubes 
having  a  value  such  that  the  plate  potential  of  the  r-f  tube  is  about  70  per  cent  that  of  the 
modulator.  This  resistor  must  be  adequately  bypassed  for  the  lowest  audio  frequency 
employed.  The  time  constant  of  the  self-biasing  circuit  in  the  grid  of  the  r-f  oscillator 
must  be  such  as  to  be  able  to  follow  the  highest  modulation  frequency  if  the  same  peak 
audio  voltage  is  to  provide  the  same  percentage  modulation  of  the  carrier. 

The  electron-coupled  oscillator  circuit  depends  for  its  frequency  stability  on  the  main- 
taining of  a  fixed  ratio  for  plate  and  screen  voltages.  This  is  best  accomplished  by  feeding 
the  screen  from  a  voltage  divider  between  plate  and  ground  of  sufficiently  low  resistance 

that  the  screen  voltage  is  in- 
/  Audio  bypass 
condenser 

ttage  dropping  resistor 


Voltage  dropping 
resistor 


FIG.  8.    Modulation  of  Impedance  Stabilized  Oscillator 


lodulator 


Direct  Coupled  Modulation  of  Electron  Coupled  Oscil- 
lator 


variant  with  screen  current. 
The  screen  must  be  bypassed 
with  a  condenser  having  low 
impedance  to  the  r-f  voltage 
but  high  impedance  to  the 
modulating  voltage.  As  in  the 
impedance-stabilized  oscillator, 
a  dropping  resistance  must  be 
employed  between  the  plates  of 
the  r-f  and  modulator  tubes  if 
100  per  cent  modulation  is  to 
be  obtained.  A  possible  varia- 
tion, of  course,  is  to  supply  the 


tubes  from  different  taps  on  a  power  supply  and  to  couple  the  modulator  to  the  r-f  oscillator 
by  means  of  a  transformer.  A  circuit  diagram  illustrating  the  former  scheme  is  shown  in 
Pig.  9,  and  the  latter  alternative  is  shown  in  Fig.  10.  This  eliminates  the  need  of  a  dropping 


11-94 


ELECTRICAL  MEASUREMENTS 


R-f  chcfee 


resistor  but  requires  a  tapped  power  supply  voltage  and  a  transformer  of  flat  response 
over  the  audio  band  width  used.  Where  extreme  freedom  from  f-m  effects  is  desired,  an 
r-f  amplifier  is  modulated  rather  than  the  r-f  oscillator.  Under  these  conditions,  the 
r-f  oscillator  works  at  constant  potential  and  is  lightly  loaded.  This  arrangement  also 
makes  the  oscillator  frequency  stable  with  changes  in  loading  of  the  output  attenuator. 
The  only  disadvantage  of  the  scheme  is  concerned  with  the  necessity  for  providing  a  tuned 
amplifier  which  is  ganged  with  the  r-f  oscillator.  The  increase  in  stability  obtained, 

however,  is  sufficiently  great  to 

Modulation  warrant  the  use  of  this  method 

with  signal  generators  of  the 
precision  type. 

Most  signal  generators  for  use 
in  testing  broadcast  receivers 
are  equipped  with  means  of 
modulating  at  400  cycles,  this 
frequency  having  been  stand- 
ardized for  receiver  testing. 
Usually,  provision  is  also  made 
for  modulation  by  means  of  an 
external  oscillator  operating  at 
any  desired  frequency  in  the 


+200  +300 


FIG.  10.    Transformer  Coupled  Modulation  of  Electron  Coupled 
Oscillator 

audio  range.  If  suitable  precautions  have  been  taken  in  choosing  the  time  constants  of 
the  grid  leak  and  condenser  biasing  system  and  of  the  plate  dropping  networks,  per  cent 
modulation  may  be  calculated  by  impressing  various  d-c  voltages  on  the  plate  of  the  r-f 
oscillator  and  measuring  the  variation  in  r-f  output  by  means  of  a  vacuum-tube  voltmeter 
or  other  method.  A  meter  which  measures  the  impressed  audio  voltage  may  then  be 
calibrated  in  terms  of  per  cent  modulation.  Various  types  of  a-c  meters  have  been  used 
for  this  purpose  such  as  thermocouple,  vacuum-tube  voltmeter,  and  rectifier-type  meters. 
Xf  other  than  sine  wave  modulation  is  externally  applied,  the  indication  of  such  meters 
must  be  corrected  accordingly. 

FREQUENCY  MODULATION.  For  use  in  visual  alignment  of  wide-band  filters  as 
well  as  for  testing  frequency-modulation  receivers,  a  frequency-modulation  generator  is 
often  required.  Several  methods  for  varying  the  radio  frequency  at  an  audio  rate  have 
been  employed.  The  simplest  of  these  consists  of  a  rotating  trimmer  condenser,  in  parallel 
with  the  oscillator  tank,  mounted  on  the  shaft  of  a  small  motor.  This  scheme  allows  wide 
frequency  variation.  It  suffers,  however,  from  several  defects,  among  them  being  troubles 
encountered  from  vibration,,  contact  troubles  if  slip  rings  are  used  for  connection,  variation 
in  amount  of  frequency  swing  as  the  center  frequency  is  varied,  and  production  of  undesired 
amplitude  modulation.  The  audio  waveform  and  frequency  of  such  a  device  are  usually 
fixed.  A  similar  scheme  makes  use  of  a  vibrating  rather  than  a  rotating  plate.  This 
allows  modulation  at  higher  audio  frequencies  and  the  frequency  may  be  more  readily 
varied.  The  obtainable  frequency  sweep  is  much  smaller,  however,  and  a  frequency 
multiplier  scheme  must  usually  be  employed  to  obtain  the  required  sweep  in  the  r-f  output 
frequency. 

A  method  which  is  more  complex  but  considerably  more  versatile  makes  use  of  a  react- 
ance tube  modulator.  The  frequency  modulation  obtainable  with  this  method  without 
large  accompanying  amplitude  modulation  is  small,  and  a  frequency  multiplier  scheme 
must  be  employed.  To  eliminate  the  necessity  of  tuning  the  multiplier  stages,  a  heterodyne, 
system  such  as  that  shown  in  Fig.  1 1  is  usually  employed. 


Fbced  r-f 
oscillator 

Frequency 

Mi 

Amplitude 
limiter 

Output  to 
attenuator 

multiplier 

Reactance 
modulator 

Var 
r-f  osc 

able    I 
:iJlator| 

Desired 
audio  waveform 

FIG.  11. 


Constant  Deviation  Frequency  Modulator 


With  such  a  system,  the  frequency  swing  of  the  output  frequency  is  independent  of  the- 
center  frequency.  A  typical  generator  of  this  type  produces  a  frequency  swing  of  over  a 
megacycle  per  second  at  center  frequencies  from  60  to  120  megacycles  per  second. 

At  microwave  frequencies,  where  velocity  modulation  tubes  are  usually  employed, 
frequency  modulation  can  readily  be  accomplished  by  applying  the  modulation  signal  to- 


MODULATION  OF  SIGNAL  GENERATOKS 


11-95 


the  reflector  of  a  reflex-type  oscillator.  At  3000  megacycles  per  second,  for  example,  a 
frequency  swing  of  approximately  30  megacycles  per  second  can  be  obtained  with  accom- 
panying amplitude  modulation  of  approximately  50  per  cent.  Frequency  swings  of  4  or  5 
megacycles  per  second  can  be  obtained  with  negligible  amplitude  modulation. 
,  PULSE  MODULATION.  For  testing  radar,  blind  landing,  and  similar  systems,  pulse- 
modulated  generators  are  required. 


Systems  using  pulse  modulation  are  usually  located  in  the  ultra-high-frequency  or 
microwave  portions  of  the  spectrum,  so  that  this  type  of  signal  generator  is  most  commonly 
found  in  these  frequency  ranges.  Requirements  for  pulse  modulation  vary  widely ._  A 
versatile  type  of  instrument  which  has  great  usefulness  provides  a  pulse  of  variable  width 
and  of  variable  delay  from  an  initiating  trigger  pulse.  Figure  12  shows  such  a  pulse 
modulator  designed  to  operate  with  a  velocity  variation  oscillator.  The  circuit  is  designed 


11-96  ELECTRICAL  MEASUREMENTS 

to  operate  either  with  an  external  trigger  voltage  applied  at  the  SYNC  IN  jack  (J-101)  or 
with  its  own  synchronizing  signal  developed  by  the  PRF  oscillator.  In  the  latter  case  the 
forward  edge  of  the  square  pulse  produced  by  the  PRF  multivibrator  V-103  is  differentiated 
in  the  grid  circuit  of  the  output  synchronizing  trigger  amplifier  V-104  so  as  to  produce  a 
short  pulse  in  the  output  of  this  tube.  This  pulse  is  then  fed  to  a  cathode  follower  output 
stage  which  delivers  a  positive  synchronizing  trigger  from  J-102.  The  output  of  either 
the  external  synchronizing  pulse  or  the  self-generated  synchronizing  pulse  is  selected  by 
switch  S103-A  and  fed  to  the  delay  multivibrator  V-106.  The  trailing  edge  of  the  pulse 
produced  in  this  multivibrator  is  differentiated  in  the  grid  circuit  of  the  pulse  amplifier 
tube  V-107.  The  negative  pulse  so  produced  drives  V-107  far  below  cutoff,  and  the  time 
required  to  reach  the  conducting  state  again  is  determined  by  the  time  constant  of  the 
differentiating  circuit  in  the  grid  of  the  pulse  amplifier  stage.  This  pulse  is  then  squared 
and  amplified  in  V-108,  V-109,  and  V-110,  and  delivered  to  the  keyer  V-lll  and  V-112. 
The  cathode  return  of  the  keyer  tube  is  connected  to  the  normal  reflector  supply  voltage 
for  the  velocity  variation  tube,  which  may  be  adjusted  to  the  correct  value  for  proper 
operation  of  the  tube  when  the  M OD-CW  switch  is  in  the  CW  position.  With  the  switch 
then  thrown  to  the  MOD  position,  and  in  the  absence  of  a  pulse,  the  reflector  potential  is 
such  as  to  preclude  oscillation.  During  the  pulse,  however,  the  reflector  voltage  is  restored 
momentarily  to  its  previous  value.  Pulsed  operation  of  the  velocity  variation  tube  is 
thus  achieved.  The  circuit  is  designed  to  operate  with  pulse  recurrence  rates  from  50  to 
5000  pps  with  pulses  delayed  from  the  trigger  pulse  from  1  microsecond  to  100  micro- 
seconds, and  having  widths  variable  from  0.5  to  30  microseconds. 

In  using  such  pulse  modulation  systems,  care  must  be  exercised  in  assuring  that  the  Q 
of  the  r-f  circuit  being  modulated  is  sufficiently  low  to  pass  the  sidebands  generated  by 
the  pulse  operation.  For  r-f  frequencies  below  100  megacycles  per  second  this  requirement 
is  difficult  to  meet  if  stable  oscillator  performance  is  to  be  obtained.  A  method  which  is 
reasonably  successful  consists  in  pulsing  an  r-f  amplifier  driven  by  the  oscillator.  The  main 
problem  encountered  here  is  to  provide  sufficient  shielding  so  that  no  r-f  output  is  present 
between  pulses.  The  degree  of  shielding  required  is  very  great  if  it  is  desired  to  test 
sensitive  receivers,  and  it  is  difficult  to  obtain  largely  because  of  coupling  due  to  internal 
electrode  capacitances  of  the  amplifier  tube.  A  second  method  consists  of  pulse-modulating 
a  higher-frequency  oscillator  and  obtaining  the  desired  frequency  by  heterodyne  methods. 
The  problem  in  applying  this  technique  is  largely  that  of  filtering  out  the  undesired  fre- 
quency components  in  the  output. 

38.  STANDARDIZATION  OF  OUTPUT  POWER 

One  of  the  most  difficult  problems  in  signal  generator  design  is  to  determine  the  r-f 
power  output  accurately.  One  method  that  has  seen  considerable  application  in  the  lower- 
frequency  ranges  is  to  measure  the  current  into  a  resistive  output  attenuator  by  means  of  a 
thermocouple.  The  quantity  of  interest  is,  of  course,  the  output  voltage  (or  power)  into 
a  specified  load  impedance.  It  is  desired,  then,  to  have  the  thermocouple  read  the  voltage 
(or  power)  at  the  input  to  the  attenuator  so  that,  the  attenuator  having  been  calibrated, 
the  output  voltage  (or  power)  will  be  known  accurately  for  all  settings  of  the  attenuator. 
Several  considerations  enter  into  the  validity  of  such  a  calibration. 

In  order  that  a  given  thermocouple  reading  will  correspond  to  a  given  voltage  or  power 
level  at  the  input  to  the  attenuator,  the  impedance  seen  looking  into  the  attenuator  from 
the  thermocouple  must  remain  constant  for  all  attenuator  settings,  assuming  that  the 
attenuator  output  is  correctly  terminated.  At  frequencies  low  enough  so  that  the  at- 
tenuator elements  can  be  considered  pure  resistances,  this  condition  can  be  met  (see 
article  39).  Difficulty  is  encountered  at  the  higher  frequencies,  however,  in  that  the 
attenuator  input  becomes  reactive  so  that  its  impedance  varies  with  frequency.  Under 
these  conditions  the  input  current  no  longer  bears  a  fixed  relationship  to  the  output  volt- 
age. At  higher  frequencies,  the  thermocouple  impedance  itself  is  also  subject  to  variation 
due  to  the  impedance  presented  to  the  heater  by  the  couple  and  its  associated  measuring 
leads.  The  design  of  an  output  filter  for  the  couple,  which  permits  a  very  low  capacitive 
admittance  to  ground  at  the  higher  frequencies,  becomes  very  difficult. 

The  difficulty  is  minimized  by  using  heater  wires  of  very  small  diameter.  Thermocouples 
of  the  separate  heater  type,  where  the  couple  is  insulated  from  the  heater  by  a  small  glass 
bead,  minimize  these  effects.  They  are,  in  turn,  sluggish  in  operation  and  less  sensitive 
than  direct-contact  types.  Correction  must  also  be  made  for  skin  effect  of  thermocouples 
used  at  high  frequencies. 

At  very  high  frequencies,  where  the  physical  distance  between  the  thermocouple  and 
tfc*  attenuator  and  between  the  thermocouple  and  the  signal  generator  oscillator  becomes 


STANDARDIZATION  OF  OUTPUT  POWER  11-97 

an  appreciable  part  of  a  wavelength,  further  trouble  is  encountered  from  reflections  from 
tne  couple  itself  as  seen  by  the  signal  generator  oscillator  and  variations  in  the  phase  of 
such  reflections  as  a  function  of  frequency  due  to  changes  in  the  electrical  lengths  involved 
as  the  frequency  changes.  For  these  reasons,  thermocouples  have  seen  their  greatest 
application  in  the  frequency  range  below  10  megacycles  per  second  although  they  may  be 
used  with  special  precautions  at  very  much  higher  frequencies. 

A  second  method  for  standardizing  output  consists  in  the  use  of  a  high-impedance 
vacuum-tube  voltmeter  at  the  input  to  the  attenuator.  It  is  usually  desirable  to  have  the 
reading  ^of  such  a  vacuum-tube  voltmeter  independent  of  modulation.  This  may  be 
accomplished  by  utilizing  an  averaging  type  voltmeter  circuit.  This  type  of  operation  is 
obtained  when  the  vacuum  tube  of  the  vacuum-tube  voltmeter  is  operated  with  large 
voltage  input  and  with  grid  leak  and  condenser  time  constant  sufficiently  small  to  allow  it 
to  follow  the  audio  modulation.  Under  these  conditions,  the  meter  reads  the  average  r-f 
voltage  applied  to  its  terminals.  Diode  voltmeters  with  large  applied  voltage  may  also 
be  used;  they  have  the  advantage  of  maintaining  calibration  with  tube  life.  It  should 
be  remembered  that  the  power  output  from  a  modulated  oscillator  is  proportional  to 
(1-1-  m2/2),  where  ra  is  the  fractional  modulation  (per  cent  modulation/ 100),  so  that  m 
must  be  known  if  it  is  desired  to  determine  the  power  delivered  to  the  attenuator  (or 
effective  voltage  at  the  attenuator  input). 

Vacuum-tube  voltmeters  may  be  used  at  frequencies  up  to  those  where  transit  time 
effects  reduce  the  rectification  efficiency  and  thus  introduce  error.  Transit  time  may  be 
minimized  by  close  electrode  spacing  and  high  applied  voltages.  Close  spacing,  however, 
introduces  large  interelectrode  capacitance,  and  this  may  itself  become  a  source  of  error. 
Interelectrode  capacitance  may  be  reduced  by  making  the  physical  size  of  the  electrodes 
small,  so  that  an  ideal  tube  for  the  purpose  would  have  very  gm^l  electrodes  spaced  close 
together.  In  addition,  it  is  necessary  to  take  special  precautions  to  minimize  the  induct- 
ance of  the  leads  to  the  tube  electrodes.  In  the  acorn-tube  types  lead  inductances  are 
sometimes  minimized  by  bringing  out  two  leads  in  parallel  from  each  electrode.  In  tubes 
such  as  the  2C40,  heavy  cylindrical  leads  are  employed.  For  acorn-type  diodes,  the  reduc- 
tion in  rectification  efficiency  is  of  the  order  of  30  per  cent  at  500  megacycles  per  second 
for  voltages  of  0.5  volt  or  less.  This  effect,  together  with  the  change  of  input  impedance 
with  frequency,  limits  the  range  of  usefulness  of  the  vacuum-tube  voltmeter  as  a  power 
output  monitor. 

The  deleterious  effects  of  transit  time  can  be  irdnimized  to  some  extent  by  substituting 
a  crystal  rectifier  for  a  vacuum  tube  in  a  voltmeter  circuit.    A  semiconductor  such  as  silicon 
in  crystalline  form  is  embedded  in  a  conducting  material  which  forms 
one  contact.    The  second  contact  is  made  through  a  tungsten  whisker. 
The  resistance  between  the  whisker  contact  and  the  semiconductor  is 
non-linear  and  may,  therefore,  be  used  to  rectify  an  impressed  alter- 
nating voltage.     Though  transit  time  in  such  a  crystal  rectifier  is 
entirely  negligible,  a  similar  effect  occurs  which  is  due  to  the  capaci- 
tance between  the  whisker  contact  and  the  crystal  which  may  be    FIG.  13.    Equivalent 
thought  of  as  shunting  the  non-linear  resistance  as  shown  in  Fig.  13.  Rectifier  Iys 

Here,  rs  represents  the  resistance  of  the  bulk  crystal  between  the 
barrier  layer  and  the  fusible  metal  in  which  the  crystal  is  embedded,  re  represents  the  non- 
linear resistance  between  the  whisker  and  the  crystal,  and  O  is  the  capacitance  between  the 
whisker  and  the  crystal  across  the  barrier  layer.  At  high  frequencies,  the  capacitance 
shunts  the  non-linear  resistance  in  such  a  way  as  to  reduce  the  rectification  efficiency, 
producing  an  effect  similar  to  that  produced  by  transit  time  in  tube  rectifiers.  With  the 
type  1N21-B  rectifier,  however,  this  effect  is  small  for  frequencies  at  least  as  high  as 
1000  megacycles  per  second.  Other  disadvantages  to  the  use  of  crystals  exist,  however, 
among  them  non-uniformity  of  characteristic  and  variation  of  impedance  among  crystals 
of  the  same  type,  and  a  change  of  characteristics  with  overload. 

For  very  high-frequency  applications,  the  variation  of  indicated  power  with  frequency 
due  to  variation  in  electrical  length  of  the  leads  to  the  power  indicator  has  led  to  the  use 
of  systems  which  divert  a  known  fraction  of  the  power  from  the  signal  generator  oscillator 
to  the  power  indicator.  Bolometer  elements  are  frequently  used  as  the  power-detecting 
devices  in  such  a  system  (see  article  11-33).  They  are  sensitive  detectors  of  r-f  power, 
having  resistance  slopes  of  several  thousand  ohms  per  watt.  For  example,  a  10-milHampere 
Littlefuse  has  a  slope  of  2800  ohms  per  watt,  while  a  5-milliampere  Littlefuse  has  a  slope 
of  5100  ohms  per  watt.  Commercially  available  Littlefuses  are  often  used  as  r-f  power 
detectors  and  are  quite  satisfactory  up  to  frequencies  of  several  hundred  megacycles  per 
second.  At  higher  frequencies,  their  geometry  is  undesirable  and  special  bolometer  ele- 
ments such  as  that  shown  in  Fig.  14  are  employed. 

The  resistance-power  curve  of  a  thermistor  is  not  accurately  linear  as  is  that  of  a  hot- 


11-98 


ELECTRICAL  MEASUREMENTS 


wire  bolometer.  This  is  of  little  importance  when  it  is  desired  to  monitor  the  r-f  power 
at  constant  level  but  becomes  important  when  it  is  desired  to  provide  an  easily  calibrated 
indicator  which  will  measure  power  over  some  range.  The  r-f  impedance  of  a  bolometer 
depends  on  its  geometry  as  well  as  on  its  resistance.  It  has  been  found  more  difficult  in 
manufacture  to  hold  the  r-f  impedance  of  thermistor  units  than  it  is  with  special  hot-wire 
bolometers. 

If  it  is  desired  to  operate  any  type  of  temperature-sensitive  power  indicator  over  wide 
ranges  of  ambient  temperature,  some  form  of  temperature-compensating  circuit  must  be 
employed.  This  problem  becomes  more  acute  as  the  sensitivity  of  the  power  indicator  is 
increased.  Temperature-sensitive  elements  such  as  those  used  in  the  power-measuring 


Attenuator 


18  gaga 
%  copper  wire 


0.0000375'V' 
platinum  wire 

FIG.  14.     Hot-wire  Bolometer 


. Mta* *WV\;     |     .  ^AAr^-VvNVj    1       -O 


FIG.  15.     Power  Monitor  Equivalent  Circuit 


circuit  may  be  employed  to  accomplish  the  required  compensation.  For  a  direct-reading 
bridge  circuit,  two  such  compensating  elements  are,  in  general,  used.  One  of  these  main- 
tains the  zero  reading,  and  the  other  adjusts  the  slope  of  the  indicator  so  as  to  maintain 
the  correct  calibration. 

In  the  use  of  power  indicators  which  absorb  a  fraction  of  the  generator  power,  care 
must  be  taken  to  insure  that  the  power  split  between  the  power  indicator  and  the  output 
circuit  is  maintained  constant  over  the  frequency  band  of  operation. 

At  the  lower  radio  frequencies  where  line  lengths  may  be  neglected  this  problem  does 
not  present  serious  difficulties.  Here,  an  arrangement  such  as  that  shown  in  Fig.  15  may 
be  used  where  the  power-sensitive  element  is  in  series  with  the  output  attenuator.  Here, 
if  the  line  length  between  the  power  detector  zp  and  the  output  attenuator  za  can  be 
neglected,  the  current  in  the  two  elements  is  common  so  that  the  power  ratio  is  simply 

Ta  "  TJM  (3) 

where  re(zp)  represents  the  real  part  of  the  power  indicator  impedance. 

At  frequencies  high  enough  so  that  the  line  length  becomes  an  appreciable  part  of  a 

wavelength,  the  current  is  no  longer  the  same  in  the  two  elements  and  the  equation  holds 

only  if  the  impedance  of  the  attenuator  is  transformed  down  the  line  to  a  point  close  to  zp. 

Then,  if  the  impedance  of  the  attenuator  is  frequency  dependent,  changes  in  both  the  real 

and  reactive  components  will  affect  the  power  split.    The  difficulty  is  overcome  only  by 

maintaining  the  attenuator  imped- 
ance close  to  the  characteristic  im- 
pedance of  the  line  over  the  fre- 
quency range. 

At  very  high  frequencies,  coaxial 
or  wave-guide  structures  are  used. 
One  method  of  monitoring  power 
from  a  cavity-type  oscillator  is 
shown  in  Fig.  16,  Here  the  power 
from  the  oscillator  is  coupled  through 

two  openings  in  the  cavity  to  wave-guide-beyond-cutoff  attenuators  which  feed  the  power 

monitoring  and  output  circuits  respectively. 
The  power  delivered  to  the  monitor  is  then: 

2 


Power  sensitive 
element  Z 


Attenuator  matching 
fmpedauce  ZA 


>ass 
condenser 


Waveguide-beyond-cutoff 
attenuators 

FIG.  16.     Power  Monitor  for  Cavity  Oscillator 


where  Ep  =  voltage  induced  in  coupling  loop  feeding  the  power  detector,  zp 
of  power  detector,  and  re(zp)  —  real  part  of  zp. 

Similarly,  the  power  delivered  to  the  attenuator  is  given  by: 


(4) 
impedance 

(5) 

where  EA  =  voltage  induced  in  coupling  loop  feeding  the  attenuator,  ZA  -  impedance  of 
attenuator  seen  looking  from  the  loop,  and  re(zA)  —  real  part  of  ZA- 
The  ratio  of  these  two  powers  is  then  given  by 


?£ 
EA 


X 


(6) 


ATTENUATOR  DESIGN  11-99 

Since  the  ratio  EP/EA  can  be  made  quite  constant  over  a  wide  frequency  band,  if  sym- 
metrical coupling  to  the  cavity  is  employed,  this  reduces  to: 


(7) 

One  way  to  make  the  right-hand  side  of  this  equation  constant  is  to  have  both  zp  and 
ZA  equal  the  real  characteristic  impedance  of  the  lines  in  which  they  are  placed. 

A  similar  situation  obtains  in  wave-guide  circuits  where  the  monitor  and  attenuator  are 
placed  in  two  arms  of  a  wavelength  T  and  To 

are  fed  from  the  third  arm  as  shown  in  bridge  *     generator 

Fig.  17.    Here  the  monitor  and  attenuator 
are  effectively  in  series.    The  power  split      eyp 


is  given  by  condenser  | ^x^ /    y*     /  >•  Output 


PA        re(zA)  FIG.  17.     Wave-guide  Power  Monitor 

This  can  also  be  made  constant  by  matching  both  attenuator  and  power  indicator  to 
the  wave  guide. 

39.  ATTETTOATOR  DESIGN 

For  frequencies  up  to  about  2  megacycles  per  second,  resistive  attenuators  using  Ayrton 
Perry  non-inductive  wire-wound  resistors  may  be  used.    In  order  to  maintain  the  input 

and  output  impedance  of  such  an  attenuator  con- 
stant as  the  attenuation  is  varied,  iterative  net- 
works of  the  T  or  a-  type  are  used.  Such  a  T 
network,  designed  to  match  without  reflection 
the  generator  impedance  rg  to  a  load  resistance 
FIG.  18.  T-type  Attenuator  TL,  is  shown  in  Fig.  IS.  If  the  generator  and  load 

are  to  be  connected  to  the  network  without  intro- 
ducing reflection,  rin  must  equal  rg  and  r0ut  must  equal  TL-  The  attenuation  constant  of 
such  a  network  is  denned  by 

•Bout  _  lout  _     -a  (Q} 

-=—  —  j—  —  e  (V) 

-Grin  -tin 

=  ln^  (10) 

-C'in  jfin 

For  given  generator  and  load  resistances  and  specified  attenuation  constant  the  elements 
of  the  network  are  given  by: 

(rt  +  TL)  tank  (f  J  +  T,  -  n  (llo) 

n _ 

(TI  +  TL)  tanh  {  -  )  -  r,  +  TL 
T2 ^ (116) 

(lie) 


2sinho: 
In  the  special  case  where  rg  =  TL  =  n>,  these  expressions  reduce  to: 

n  =  r2  =  r0  tanh  |  (12o) 

rs  =  -rV  (126) 

sinh  a. 

The  -unsymmetrical  case  is  of  interest  in  signal  generators  for  the  broadcast  band  since 
it  is  sometimes  desired  to  make  the  output  impedance  sufficiently  low  that  the  voltage 
developed  across  it  is  independent  of  the  impedance  connected  to  the  output  terminals. 
The  attenuation  of  the  network  may  be  varied  and  constant  input  and  output  impedances 
maintained  if  the  resistances  are  kept  in  the  ratio  denned  by  the  equations. 

In  practice,  accuracy  requirements  and  difficulty  in  shielding  output  from  input  make 
it  undesirable  to  utilize  a  single  T  or  x  section  for  covering  large  ranges  of  attenuation. 


11-100 


For  the*  reasons,  a 


ELECTRICAL  MEASUREMENTS 

attenuator  is  commonly  employed.     A  useful  form  of 
19     If  K  =  the  ratio  by  which  the  voltage  across  rL 
;,  the  resistances  n  and  r2  are  given  by : 

(13a) 


TL  (K  -  1) 


r2  =  rz, 


(136) 


The  input  resistance  rm  is  independent  of  the  position  of  the  tap  switch  and  is  given  by: 


(U) 


where 


+  4r2)  + 


Between  TL  and  the  step  attenuator,  it  is  usually  desirable  to  place  a  variable  T  pad 
attenuator  of  the  type  previously  described  in  order  to  obtain  a  fine  control.  TL  may  be 
t^e  external  load  resistance  into  which  the  generator  is  designed  to  feed,  or  it  may  be  made 


Resistive  center  conductor 


ru  Output 


FIG.  19.     Ladder-type  Attenuator  with 
Constant  Input  Impedance 


V 

Resistive  disks 
FIG.  20. 


Coaxial  IT  Sec- 
tion 


ResistJve  matching  transformers 


low  in  value  and  incorporated  into  the  signal  generator  in  which  case  the  actual  load 
impedance  must  be  large  compared  to  rL  for  proper  operation.  For  satisfactory  operation, 
each  section  of  the  attenuator  must  be  separately  shielded  and  a  separately  shielded  low- 
capacity  switch  must  be  employed.  ,  _  .,  .  .  ,  ,.  . 

For  high-frequency  applications,  T  or  TT  sections  may  be  built  in  coaxial  line  iorm  as 
shown  in  Fig.  20.    Units  of  this  type  may  be  used  up  to  the  frequency  where  the  length  of 

line  becomes  an  appreciable  part  of  a  wavelength,  and 
they  have  seen  successful  operation  up  to  frequencies  as 
3  Dielectric  high  as  1000  megacycles  per  second.  The  resistive  ele- 
ments may  conveniently  be  made  of  glass  rods  and  disks 
coated  with  thin  metallic  films.  A  decade  attenuator  may 
be  made  by  mounting  several  such  elements  in  a  turret 
arrangement. 

At  microwave  frequencies,  attenuators  consisting  of 
resistive  sections  of  coaxial  line  equipped  with  resistive  matching  transformers  have  been 
successfully  employed.  A  diagram  of  such  a  unit  is  shown  in  Fig.  21. 

The  band  width  of  such  devices  is  limited  by  the  change  in  attenuation  with  frequency 
and  by  the  frequency  sensitivity  of  the  resistive  matching  transformers. 

A  type  of  attenuator  which  has  seen  increasing  use  in  signal  generator  applications  is 
shown  in  Fig.  22.    This  type  is  known  as  a  wave-guide-beyond-cutoff  attenuator  and  makes 


Resistive  attenuating  section 
FIG.  21.    Microwave  Attenuator 


•*-L~* 

^   ^.<^v^Wv^ 

t 
D 
X    lA 

rpW/////////////V    I         \ 

1    I 

V 

Coupling  loops 

Loop  coupling  attenuation  =  32.0  ^1  -  (  '  2     )   db  Per  diameter  displacement  of  loops. 

Disk  coupling  attenuation  =  41.8  \1  ~  (  — -« —  )    db  per  diameter  displacement  of  loops. 
FIG.  22.     Waveguide-beyond-cutoff  Attenuator 

use  of  the  fact  that,  for  diameters  below  a  critical  value  which  depends  on  the  frequency 
and  mode  of  propagation,  waves  in  a  hollow  tube  suffer  no  phase  displacement  but  are 
damped  exponentially  in  amplitude.  Inductive  coupling  between  loops  and  capacitive 
coupling  between  disks  in  such  a  hollow  tube  may  be  thought  of  as  taking  place  in  this 


SHIELDING  PROBLEMS  11-101 

way.  The  attenuation  in  decibels  is  then  linear  with  displacement  and  follows  the  law 
given  in  the  figure  for  the  two  cases  of  loop  and  disk  coupling  respectively.  The  loop- 
coupled  attenuator  has  the  advantage  that  its  rate  of  attenuation  is  lower  than  that  of 
any  other  mode  and  is  thus  less  susceptible  to  errors  due  to  undesired  coupling  to  other 
modes,  since  these  die  away  more  rapidly  and  become  unimportant  at  any  but  the  lowest 
attenuator  settings. 

At  low  frequencies,  a  coil  of  several  turns  is  usually  employed  in  order  to  reduce  the 
minimum  insertion  loss.  For  low-frequency  applications  where  the  skin  depth  is  appre- 
ciable this  effect  must  be  taken  into  account  since  it  modifies  the  effective  diameter.  This 
may  be  done  by  adding  2p  to  the  measured  diameter,  where  p  is  the  skin  depth.  For 
copper,  p  is  given  by: 

6.6 
P  =  —F  X  1CT3     cm  (15) 

V/ 

where  /  =  megacycles  per  second. 

The  input  and^output  impedance  of  this  type  of  attenuator  is  reactive,  and  it  is  necessary 
to  provide  resistive  pads  to  match  the  generator  and  load.  These  pads  contribute  to  the 
initial  insertion  loss,  which  is  of  the  order  of  25  db  in  practical  designs.  The  high  initial 
insertion  loss  limits  the  use  of  this  type  of  attenuator,  although  it  is  satisfactory  for  many 
signal  generator  applications  where  the  power  level  of  an  oscil- 
lator must  be  reduced  to  the  noise  level  of  a  sensitive  receiver.  vwvv  o 

The  power  delivered  to  a  load  impedance  from  any  attenuator        /J 

system  can  be  analyzed  in  terms  of  Thevenin's  theorem  as  shown     E  ^>  terminals 

in  Fig.  23.    Here  zg  is  the  impedance  seen  looking  into  the  out- 
put  terminals  and  E  is  the  open-circuit  voltage  measured  at  these 


terminals.    At  low  frequencies,  zg  is  often  made  small  compared    ^  J3  A 

to  the  load  impedances  with  which  the  generator  will  operate  so 

that  the  output  voltage  is  essentially  independent  of  ZL~    A  dummy  antenna  which  simu- 

lates the  impedance  of  an  actual  receiving  antenna  is  then  placed  between  the  signal 

generator  terminals  and  the  receiver  input. 

At  higher  frequencies,  line  length  between  the  signal  generator  and  load  cause  consider- 
able difficulty  so  that  a  better  scheme  is  to  make  zt  equal  to  the  characteristic  impedance 
of  the  line  which  will  be  used.  The  load  impedance  is  then  also  matched  to  the  line  so 
that  the  voltage  at  the  load  is  E/2.  If  the  receiver  to  be  tested  does  not  present  an  input 
impedance  equal  to  the  characteristic  impedance  of  the  line,  it  may  be  transformed  to  this 
impedance  by  means  of  a  suitable  transformer.  If  this  is  not  done,  an  error  will  be  intro- 
duced which  can  be  calculated  by  means  of  the  transmission-Hne  equations, 

40.  SHIELDING  PROBLEMS 

There  is  usually  a  difference  in  level  of  the  order  of  100  db  between  the  signal  generator 
oscillator  power  and  the  attenuator  output  power  when  the  generator  is  used  for  receiver 
testing.  This  makes  necessary  extreme  precautions  in  the  matter  of  shielding.  The  depth 
of  penetration  in  a  metal  is  defined  as  that  depth  to  which  the  electric  or  magnetic  field 
falls  off  to  1/8  of  its  value  at  the  surface,  where  e  =  2.718.  An  attenuation  of  100  db 
requires  11.5  skin  depths.  At  a  frequency  of  1  megacycle  per  second,  this  corresponds  to 
0.076  cm  in  copper.  Thus,  except  for  very  low  frequencies,  the  thickness  of  the  metal 
required  is  determined  by  mechanical  rather  than  electrical  considerations.  The  real 
problem  is  concerned  with  the  bringing  in  of  d-c  leads  and  control  shafts  to  the  oscillator 
and  with  the  necessity  of  providing  removable  covers  for  the  shield  case.  Separate  shield- 
ing boxes  should  be  provided  for;the  main  units  in  the  signal  generator  such  as  the  oscil- 
lator, the  power  measurer,  and  the  attenuator  system.  These  individual  shields  are  then 
enclosed  in  an  overall  shield  and  are  preferably  grounded  to  it  at  a  single  point  to  eliminate 
circulating  currents  which  can  induce  voltages  in  lead  filters  or  other  output  connections. 
Gaskets  made  of  compressed  woven  metal  are  useful  in  preventing  leakage  from  removable 
covers. 

For  low-frequency  lead  filters,  conventional  XC-type  filters  may  be  used.  At  higher 
frequencies,  particular  attention  must  be  paid  to  the  condensers  used  since  these  may, 
in  fact,  become  inductive.  Condensers  of  the  button  type  are  particularly  useful  for  this 
purpose. 

At  frequencies  above  2000  megacycles  per  second,  lossy  filters  may  be  employed  to 
advantage.  A  filter  consisting  of  a  coaxial  line  having  powdered  iron  between  the  con- 
ductors represents  one  such  type  of  filter  which  has  seen  considerable  application.  At 
3000  megacycles  per  second,  such  a  filter,  4  in.  in  length,  can  be  made  to  have  an  attenua- 
tion of  more  than  100  db, 


11-102  ELECTRICAL  MEASUREMENTS 

Control  shafts  may  conveniently  be  brought  into  shielded  generators  by  utilizing  the 
wave-guide-beyond-cutoff  principle  and  employing  Bakelite  shafts  feeding  through  metal 
tubes  which  are  soldered  to  the  shielding  box.  A  tube  having  an  inside  diameter  of  1/4  in. 
will,  for  instance,  have  an  attenuation  of  about  100  db  for  each  3/4  in.  of  length  for  frequen- 
cies up  to  15,000  megacycles  per  second.  Another  method  of  feeding  shafts  through  a 
shielded  partition  is  by  means  of  a  flexible  diaphragm  type  of  coupler  which  solders  to  the 
shield  box  and  which  transforms  a  nutating  motion  into  a  rotary  one  in  a  manner  which 
allows  metallic  continuity  through  the  coupling. 

41.  POWER  MEASUREMENT 

Power  measurers  are  of  two  types,  one  which  samples  the  power  in  a  transmission  line 
between  generator  and  load  and  which  absorbs  only  a  small  fraction  of  the  power  being 
measured,  and  a  second  type  which  absorbs  all  the  power  being  measured,  in  general  con- 
verting it  to  heat.  A  vacuum-tube  voltmeter  of  impedance  high  compared  to  the  generator 
or  load  is  an  example  of  the  former  type;  a  bolometer  or  crystal  power  measurer  terminating 
a  transmission  line  is  an  example  of  the  latter  type. 

All  the  types  of  power  measurers  discussed  under  the  section  on  signal  generators  may 
be  extended  to  measure  higher  powers  through  the  use  of  attenuators  appropriate  to  the 
frequency  range  involved.  Bolometer-type  power  measures  of  the  hot-wire  or  thermistor 
type  are  particularly  adaptable  to  this  use  since  they  may  conveniently  be  matched  into 
coaxial  lin.es.  Again,  broad-band  designs  may  be  evolved  up  to  frequencies  where  the 
length  of  the  bolometer  element  becomes  an  appreciable  part  of  a  wavelength.  By  means 
of  special  matching  techniques,  broad-band  designs  have  been  made  up  to  frequencies  of 
10,000  megacycles  per  second. 

Attenuators  of  resistive  center  conductor  coaxial  line  using  the  thin  metallic  film  on 
glass  techniques  can  measure  powers  as  great  as  a  few  watts.  For  higher  power,  lossy 
attenuators  which  can  be  matched  over  a  considerable  frequency  range  can  be  made  by 
filling  the  space  between  conductors  in  a  coaxial  line  with  a  lossy  compound  such  as  may 
be  made  with  graphite  or  iron  powder  and  a  suitable  cement  binder.  With  proper  propor- 
tions of  lossy  material  to  binder  the  loss  per  unit  length  can  be  adjusted  to  permit  the  power 
absorbed  to  be  radiated  adequately  from  the  length  of  line  used. 

BIBLIOGRAPHY 

Terman,  F.  E.,  Radio  Engineering,  McGraw-Hill  Book  Co.,  1938. 

Terman,  F.  E.,  Measurements  in  Radio  Engineering,  McGraw-Hill  Book  Co.,  1939. 

Sarbacher,  R.  I.,  and  Edson,  W.  A.,  Hyper  and  Ultra-high  Frequency  Engineering,  John  Wiley  and  Sons, 

McArthur,  E.  D.,  and  Spitzer,  E.  E.,  Vacuum  Tubes  as  High  Frequency  Oscillators,  Proc.  I.R.E.,  Vol. 

19,  No.  11,  1971  (November  1931). 
Seeley,  S.  W.,  and  Anderson,  E.  I.,  U.H.F.  Oscillator  Frequency  Stability,  R.C.A.  Rev.,  Vol.  5,  77 

(July  1940). 

Harrison,  A.  E.,  Klystron  Technical  Manual,  Sperry  Gyroscope  Co.,  1944. 

Karplus,  E.,  The  Butterfly  Circuit,  General  Radio  Experimenter,  Vol.  19,  No.  5  (October  1944). 
Mayer,  H.  F.,  Visual  Alignment  Generator,  Electronics,  Vol.  13,  No.  4,  39  (April  1940). 
Crosley,  M.  G.,  Reactance  Tube  Frequency  Modulator,  Q.S.T.,  June,  1940. 
Sheaffer,  C.  F.,  Frequency  Modulator,  Proc.  I.R.E.,  Vol.  28,  No.  2,  66  (February  1940). 
Peterson,  A.,  Vacuum  Tube  and  Crystal  Rectifiers  as  Galvanometers  and  Voltmeters  at  Ultra-high 

Frequencies,  General  Radio  Experimeter,  Vol.  19,  No.  12  (May  1945). 

Becker,  J.  A.,  Green,  C.  B.,  and  Pearson,  G.  L.,  Properties  and  Uses  of  Thermistors — Thermally  Sensi- 
tive Resistors,  Electrical  Engineering,  Vol.  65,  No.  11,  Transactions,  p.  711  (November  1946). 
Smith,  M.  T.,  An  Improved  Ultra-high  Frequency  Signal  Generator,  General  Radio  Experimenter,  Vol. 

15,  No.  8  (February  1941). 
Scott,   H.  H.,  Progress  in  Signal  Generator  Design,   General  Radio  Experimenter,  Vol.    17,   No.   6 

(November  1942). 
Harnett,  D.  E.,  and  Case,  N.  P.,  The  Design  and  Testing  of  Multirange  Receivers,  Proc.  I.R.E.,  Vol. 

23,  No.  6,  578  (June  1935). 
McElroy,  P.  K.,  Designing  Resistive  Attenuating  Networks,  Proc,  I.R.E.,  Vol.  28,  No.  2,  66  (February 

Peterson,  A.,  Output  Systems  of  Signal  Generators,  General  Radio  Experimenter,  Vol.  21,  No.  1  (June 
Shive,  S.  L.f  Effectiveness  of  Conduit  as  R.F.  Shielding,  Electronics,  Vol.  19,  No.  2,  160  (February  1946). 


SECTION  12 
ACOUSTICS 


THE  SENSE  OF  HEARING 

By  JOHN  C.  STEINBERG  AND 
^T.  W.  A.  MXJNSON  PAGE 

1.  Description  of  the  Ear 02 

2.  Sensitivity  of  the  Ear 05 

3.  Differential  Sensitivity 09 

4.  Masking  Effects  of  Sounds 11 

5.  Loudness  of  Sounds 11 

6.  The  Pitch  of  Steady  Sounds 16 

7.  Localization  of  Sounds 18 

SPEECH  AND  MUSIC 
BY  JOHN  C.  STEINBERG  AND  W.  A.  MTJNSON 

8.  Description  of  Speech  Organs 19 

9.  Production  of  Speech 19 

10.  Speech  Power 22 

11.  Powers    Produced    by    Musical    Instru- 

ments       24 

12.  Tests  of  Speech  and  Music  Transmission     27 

EFFECTS  OF  DISTORTION  ON  SPEECH 

AND  MUSIC 
BY  JOHN  C.  STEINBERG  AND  W.  A.  MTJNSON 

13.  Effect  of  Frequency  Distortion 30 

14.  Articulation  Tests 31 

15.  Auditory  Perspective 39 

ACOUSTIC  PROPERTIES  OF  ROOMS 

By  VEEN  O.  KNTJDSEN 

16.  Requirements  for  Good  Acoustics 40 

17.  Geometric  and  Wave  Acoustics 41 


AET.  PAGE 

IS.  Growth  and  Decay  of  Sound  in  Rooms 

— General  Considerations 42 

19.  Reverberation  Equations 43 

20.  Room  Resonance 45 

21.  Reverberation  at  Different  Frequencies.  47 

22.  The  Measurement  of  Reverberation  and 

Absorption  Coefficients 48 

23.  Coefficients  of  Sound  Absorption 50 

24.  Practical  Considerations   of   Sound-ab- 

sorptive Materials 57 

SOUND  INSULATION 
BY  VEBN  O.  EJSFUDSEN 

25.  Noise  Measurements 57 

26.  Acceptable    Noise    Levels   in    Different 

Buildings 58 

27.  Fundamental  Principles  of  Sound  Insu- 

lation      60 

28.  Coefficients  of  Sound  Transmission 64 

29.  Practical  Considerations  in  the  Selection 

of  Materials  and  Types  of  Structure 
for  Insulation  in  Buildings 67 

30.  Calculation    of   Insulation    in    Building 

Design 69 

ACOUSTIC  DESIGN  OF  AUDITORIUMS 
BY  VEBN  O.  KNTJDSEN 

31.  The  Hearing  of  Speech  in  Auditoriums.     69 

32.  Music  Rooms 74 

33.  Practical  Procedure  for  Obtaining  Good 

Acoustics  in  Buildings 76 


12-01 


ACOUSTICS 


THE  SENSE  OF  HEARING 

By  John  C.  Steinberg  and  W.  A.  Munson 

In  the  design  of  sound  transmission  and  reproduction  systems,  consideration  should  be 
given  to  the  physical  and  physiological  properties  of  the  voice  and  ear,  and  to  the  charac- 
teristics and  psychophysiological  effect  of  the  different  types  of  sounds  that  the  systems 
•are  called  upon  to  transmit  and  reproduce. 

1.  DESCRIPTION  OF  THE  EAR 

The  hearing  mechanism  is  usually  divided  into  three  parts,  the  outer  ear,  the  middle  ear, 
and  the  inner  ear.  A  cross-section  of  the  ear  is  shown  schematically  in  Fig.  1. 


PIG.  1.     Semi-schematic  Section  of  Left  Ear.    P,  pinna;  E,  ear  canal;  D,  eardrum;  A,  auditory  ossicles; 
window;  U,  Eustachlan  tube;  S,  semicircular  canal;  N,  auditory  nerve; 
C,  cochlea. 


O,  oval  window;  R,  round  wind 


THE  OUTER  EAR.  The  ear  canal  (E)  has  a  length  of  about  2.5  cm,  a  volume  of 
1,0  cu  cm,  and  an  area  at  the  opening  of  0.3  to  0.5  sq  cm.  The  eardrum  (D),  stretched 
across  the  inner  end  of  the  canal,  has  a  horizontal  diameter  of  1.0  cm,  a  vertical  diameter 
of  0.8  cm,  and  an  area  of  0.6  sq  cm.  All  these  dimensions  vary  somewhat  for  different 
individuals. 

THE  MIDDLE  EAR.  The  middle  ear  is  separated  from  the  outer  ear  canal  by  the 
eardrum.  Motion  of  the  eardrum  is  transmitted  across  the  middle-ear  cavity  to  the  inner 
ear  by  means  of  a  system  of  levers  (A)  called  the  auditory  ossicles.  The  ossicles  consist 
of  three  small  bones,  the  malleus  (hammer) ,  attached  to  the  eardrum,  the  stapes  (stirrup) , 
attached  to  the  oval  window  '0)  of  the  inner  ear,  and  the  incus  (anvil) ,  which  is  the  con- 
necting link  between  the  malleus  and  stapes.  The  weights  of  the  three  bones  are :  hammer, 
O.023  gram;  anvil,  0.025  gram;  and  stirrup,  0.003  gram.  The  middle-ear  cavity  is  normally 
filled  with  ah*  maintained  at  atmospheric  pressure  by  virtue  of  the  Eustachian  tube  (IT) 
-which  leads  to  the  back  part  of  the  throat. 

12-02 


DESCRIPTION  OF  THE  EAR 


12-03 


THE  INNER  EAR.  The  inner  ear,  which  serves  a  dual  purpose,  is  a  complex  labyrinth 
of  liquid-tilled  passageways  imbedded  in  the  temporal  bone  adjacent  to  the  middle-ear 
cavity.  There-are  three  semicircular  canals,  oriented  so  as  to  lie  in  three  mutually  per- 
pendicular planes,  and  these  ducts  contain  the  nerve  endings  concerned  with  the  mainte- 
nance of  body  equilibrium.  Only  one  canal  (5)  is  shown  in  Fig.  1.  The  cochlea  (O  is  a 
spiral  duct  containing  the  auditory  nerve  endings.  In  Fig.  1,  it  has  been  drawn  on  an 
enlarged  scale  relative  to  the  middle  and  outer  ears,  and  the  bone  in  which  the  cochlea  is 
imbedded  is  not  shown.  The  mean  diameter  of  a  semicircular  canal  is  about  1.0  cm.  The 
cochlear  spiral  has  a  mean  diameter  of  about  0.6  cm  for  the  large  turn  at  the  base.  As 
may  be  seen  in  the  sectional  view  of  Fig.  2,  the  cochlea  is  divided  by  flexible  membranes 
into  three  canals,  scala  vestibuli  (7),  scala  media  (J/),  and  scala  tympani  (T). 

The  auditory  nerve  fibers  running  between  the  cochlea  and  the  brain  number  about 
29,000  They  terminate  with  complex  interconnections  along  the  basilar  membrane  (B), 


FIG.  2.     Semi-diagrammatic  Section  of  the  Cochlea.     (From  Gray's  Anatomy.)    B,  basilar  membrane; 
V,  scala  vestibuli;  M,  scala  media;  T,  scala  tympani;  -V,  cochlear  nerve;  L,  lamina  spiralis  ossea.- 

which  extends  from  the  bony  ledge  (L}  to  the  opposite  wall  of  the  cochlea,  separating 
scala  tympani  and  scala  media.  The  nerve  endings  are  associated  with  tiny  hair  cells 
which  protrude  into  the  liquid  filling  the  small  triangular  scala  media.  Scala  vestibuli 
and  scala  tympani  are  interconnected  at  the  apex  of  the  spiral  by  a  small  opening  called 
the  helicotrema.  At  the  base  of  the  spiral,  scala  tympani  opens  into  the  middle-ear  cavity 
through  the  round  window  (#),  shown  in  Fig.  1,  but  the  liquid  content  is  retained  by  a 
flexible  membrane  stretched  across  the  opening.  The  oval  window  (0),  between  the 
middle  ear  and  scala  vestibuli,  is  closed  by  the  foot  of  the  stapes  and  connecting  ligaments- 
Excitation  of  the  Auditory  Nerves.  A  sound  wave  in  the  outer  ear  canal  produces 
motion  of  the  eardrum  and  associated  ossicles  which,  in  turn,  agitate  the  liquid  in  scala 
vestibuli  through  the  oval  window.  When  the  stimulus  is  a  pure  tone  at  a  low  level,  this 
results  in  an  excitation  of  a  small  number  of  the  nerve  endings  enclosed  by  the  adjacent 
scala  media.  When  the  frequency  of  the  tone  is  changed,  the  excitation  moves  along  the 
basilar  membrane,  engaging  a  different  set  of  nerve  endings.  In  this  way  the  ear  differ- 
entiates between  tones  of  different  pitch.  When  the  intensity  of  the  tone  is  increased,  the 
excitation  spreads  out,  and  additional  nerve  endings  are  stimulated,  but  those  comprising 
the  initial  group  probably  receive  the  maximum  stimulation.  The  positions  of  the  stimu- 
lation maxima  are  shown  in  Fig.  3  as  a  function  of  frequency. 

At  low  frequencies  the  stimulation  maxima  are  very  broad  and  the  positions  are  poorly 
defined.  It  may  be  that  the  pitch  of  low-frequency  tones  is  sensed  by  the  stimulation 
frequency  as  well  as  by  its  position. 

NERVE  CONDUCTION.  Physiological  research  indicates  that  a  nerve  conducts  on 
an  "all-or-none"  basis.  The  magnitude  of  the  nerve  impulse  and  the  rate  of  propagation 
are  independent  of  the  manner  of  excitation.  After  an  impulse  has  been  transmitted  by 
a  nerve,  it  must  recover  its  conducting  properties  before  it  is  capable  of  being  excited 
again.  This  recovery  period  lasts  for  a  time  interval  of  1  to  3  milliseconds  and  varies 


12-04 


ACOUSTICS 


somewhat  with  the  intensity  of  the  stimulus.    Thus  a  nerve  fiber  is  unable  to  transmit  the 

wave  form  of  tne  exciting 
stimulus.  Excitation  of  the 
auditory  nerves  and  conduc- 
tion of  neural  pulses  to  the 
brain  have  been  studied  by 
placing  a  small  electrode  on  a 
single  auditory  nerve  tract  of 
anesthetized  animals  and  re- 
cording the  potential  gener- 
ated. It  was  found  that  in- 
creasing the  intensity  of  a 


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pure  tone  increases  the  rate 
at  which  pulses  are  trans- 
mitted. Typical  results  are 
shown  in  Fig.  4.  Since  a  max- 
imum rate  is  attained  when 
the  sound  level  is  increased 
about  50  db,  other  fibers  must 
be  involved  in  covering  the 
full  intensity  range  of  the  ear. 
It  is  believed  that  the  loud- 
ness  of  a  sound  is  related  to 
the  rate  at  which  neural  im- 
pulses reach  the  brain  from 
all  portions  of  the  basilar 
membrane. 

ACOUSTIC  IMPEDANCE 
OF  THE  EAR.  The  acous- 
tic impedance  which  an  ear 
presents  to  a  telephone  re- 
ceiver has  been  measured  at 


Distance  from  Hellcotrema  In  MM 

FIG.  3.    Relation  between  Frequency  of  a  Tone  and  the  Position 

of    Nerve    Endings    on    the   Basilar  Membrane  Which   Receive 

Maximum  Stimulation.    (Steinberg.) 

the  aperture  of  a  receiver  cap.  The  results  depend  to  a  large  extent  upon  the  way  the 
cap  fits  the  ear.  Typical  values  are  shown  in  Fig.  5  for  the  case  when  the  cap  is  sealed  to 
the  ear  and  for  the  case  when  an  air 
leak  is  present  between  the  cap  and  the 
ear.  These  data  are  useful  in  receiver 
design,  and  Inglis,  Gray,  and  Jenkins 
describe  an  artificial  ear  which  they 
use  for  the  measurement  of  telephone 
receivers.  It  consists  of  a  conduit  hav- 
ing the  approximate  dimensions  and 
impedance  of  a  typical  ear  canal  over 
the  important  frequency  range.  The 
exposed  end  of  the  conduit  is  fitted 
with  a  tapered  rubber  seating  surface 
on  which  the  receiver  is  placed.  The 
other  end  of  the  conduit  is  terminated 
by  an  acoustic  network  and  by  the 
diaphragm  of  a  small  condenser  trans- 
mitter. The  pressures  developed  at 
the  transmitter  diaphragm  for  a  given 
voltage  on  a  receiver  placed  on  the 
artificial  ear  are  closely  equal  to  the 
pressures  that  would  be  produced  by 
the  receiver  at  the  drum  of  a  typical 
human  ear. 

NATURAL  FREQUENCY  AND 
DAMPING  CONSTANT  OF  THE 
EAR.  The  work  of  Bek&sy  indicates 
that  the  frequency  with  which  the 
eardrum  and  ossicles  vibrate  when 
suddenly  released  from  a  displaced 
position  is  of  the  order  of  1200  to  1500 


10 


60 


70 


cycles.     The  vibrations  decay  at  a 


20          30          40          50 
Relative  Intensity  Level,  db 

FIG.  4.     Relation  between  Sound  Intensity  and  Nerve 
Discharge  Rate  for  a  1050-cycle  Tone.    (Galambos  and 


Davis.) 


SENSITIVITY  OP  THE  EAR 


124)5 


rate  of  1200  db  per  sec.  Davis  and  associates'  experiments  on  the  electrical  response  of 
the  ears  of  cats  indicate  a  natural  frequency  from  600  to  1000  cycles  and  a  decay  rate  of 
1000  db  per  sec.  Their  work  also  indicates  that  the  time  lag  between  the  sudden  applica- 
tion of  a  sound  wave  on  the  eardrum  and  the  transmission  of  the  impulse  along  the  auditory 
nerve  is  of  the  order  of  2  to  3  milliseconds.  On  the  other  hand,  the  sensory  build-up  time, 
e.g.,  the  time  needed  for  a  suddenly  applied  steady  wave  to  build  up  to  a  steady  loudness, 
is  of  the  order  of  0.2  to  0.25  sec. 

200 


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—150 
—200 
i°50 

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1.  Re 
2.  Re 

Distance  of  Typical  Male  Human  Ear,  Sealed, 
actance  of  Typical  Male  Human  Ear,  Sealed, 
slstance  of  Typical  Male  Human  Ear,  wlfh  Leafc 
actance  of  Typical  Male  Human  Ear,  with  Leafc 

—300 
1( 

3.  Re 
4.  Re 

DO                                                          500                    1000                                                        5CX 

Frequency  In  Cycles  pej  Second 

FlG.  5.     Acoustic  Impedance  of  Ears  as  Viewed  through  Aperture  of  Receiver  Cap.     (Inglis, 

and  Jenkins.) 


2.  SENSITIVITY  OF  THE  EAR 

Ear  sensitivity  is  concerned  with  the  least  intense  sound  that  can  be  heard.  Such  a 
sound  is  said  to  be  at  the  "threshold  of  hearing."  Two  classes  of  ear-sensitivity  deter- 
minations have  been  reported,  "minimum  audible  pressure"  (M.A.P.)  and  "minimum 
audible  field"  (M.A.F.).  M.A.P.  is  the  just-audible  sound  pressure  measured  near  the 
observer's  eardrum.  M.A.F.  is  the  free  field  sound  intensity  of  a  plane  progressive  wave 
that  is  just  audible  to  an  observer  facing  the  source  and  listening  binauraily:  The  sound 
intensity  is  measured  before  the  observer's  head  is  inserted  in  the  field.  It  is  convenient 
to  express  minimum  audible  values  in  decibels  from  an  arbitrary  reference.  Reference 
intensity  has  been  chosen  as  10~16  watt  per  square  centimeter.  Corresponding  reference 
pressure  at  20  deg  cent  and  76  cm  of  Hg  is  2  X  10  ~4  dyne  per  square  centimeter.  In- 
tensity and  pressure  levels  are  the  number  of  decibels  from  the  above  references,  respec- 
tively. Table  1  gives  minimum  audible  values  derived  by  Sivian  and  White  from  their 
own  work  and  that  of  others  when  a  small  selected  group  of  observers  with  excellent  hear- 
ing is  used. 

Owing  to  the  diffraction  effect  of  the  head  and  to  the  difference  between  one-ear  and 
two-ear  listening,  it  would  be  expected  that  M.A.P.  and  M.AJT.  would  differ  considerably 
at  high  frequencies  but  not  at  low  frequencies.  Various  possible  causes  of  the  differences 
shown  in  Table  1  at  the  low  frequencies  are  discussed  by  Sivian  and  White,  with  the  con- 
clusion that,  at  the  present  time,  a  satisfactory  explanation  is  not  evident. 


12-06 


ACOUSTICS 


Table  1*    Monaural  Minimum  Audible  Pressure  Levels  in  the  Ear  Canal  and  Binaural 
Minimum  Audible  Sound  Field  Intensity  Levels 


Frequency 

60 

100 

200 

500 

1,000 

2,000 

5,000 

10,000 

15,000 

M.A.P.,  db 
(Monaural)  .  .  . 
M.A.F.,  db 

(BiriAiirp.1)  .  ,  ,  , 

59 
45 

46 
33 

29 
19 

14 
8 

8 
3 

5 
-6 

12 
_3 

26 
11 

44 
22 

BINAURAL  VS.  MONAURAL.  Minimum  audible  values  obtained  with  observers 
listening  binaurally  (two  ears)  are  smaller  on  the  average  than  those  obtained  with  ob- 
servers listening  monaurally  (one  ear).  According  to  Fletcher  and  Munson,  the  differ- 
ence appears  to  be  accounted  for  by  inequalities  in  the  sensitivities  of  the  two  ears  of  an 
observer.  They  report  that  the  binaural  sensitivity  is  practically  equal  to  that  of  the 
better  ear,  and  that  monaural  sensitivity  is  equal  to  that  obtained  by  averaging  the  sensi- 
tivities of  both  ears.  Figure  6  gives  the  differences  between  the  sensitivity  of  the  better 


Acuity  Difference,  db 
^0  to  4*.  en  to  a 

' 

^ 

c 

->{' 

n 

^ 

." 

' 

1 

> 

0      50 

100        200             500      1000      2000          5000     10000    20000 
Frequency  In  Cycles  per  Second 

FIG.  6.    Difference  between  tne  Sensitivity  of  the  Better  Ear  and  the  Average  of  Both  Ears.    (Fletcher 

and  Munson.) 

ear  and  the  average  of  both  ears,  based  on  the  audiograms  of  80  persons  of  normal  hearing. 
It  gives  a  means  of  converting  binaural  M.A.F.  into  monaural  M.A.F. 

VARIATION  WITH  DIRECTION.  The  values  of  M.A.F.  given  in  Table  1  are  for 
an  observer  facing  the  source,  i.e.,  for  a  progressive  wave  whose  wave  front  is  vertical  and 
whose  direction  is  normal  to  a  line  joining  the  observers  ears.  Owing  to  diffraction  effects 
of  the  head,  the  ear  is  directive  and  generally  hears  best  when  the  open  ear  is  turned  toward 
the  source.  The  directivity  of  hearing  as  reported  by  Sivian  and  White  is  given  in  Fig.  7. 
Directivity  is  expressed  as  the  variation  in  monaural  M.A.F.  with  the  azimuth  of  the 
vertical  wave  front:  0°  corresponds  to  the  observer  facing  the  source;  +90°  corresponds  to 
the  open  ear  toward  the  source.  The  variation  is  expressed  in  decibels  from  the  M.A.F. 
at  0°.  A  positive  value  of  directivity  for  any  angle  means  that  the  ear  is  more  sensitive 
at  that  angle  than  at  0°.  By  means  of  the  directivity  data,  Sivian  and  White  have  com- 
puted the  sensitivity  of  the  ear  for  the  case  when  an  observer  is  exposed  to  a  diffuse  sound 
field,  such  that  sound  waves  of  equal  amplitudes  and  random  phase  angles  are  equally 
probable  from  all  angles.  Their  results  are  shown  in  Table  2. 

Table  2.     Binaural  Mini -mum  Audible  Sound  Field  (Random  Horizontal  Incidence). 

(Sivian  and  WMte) 


Frequency 

60 

100 

200 

500 

1,000 

2,000 

5,000 

10,000 

15,000 

Binaural  M.A.F.  T  db 
(random  horizontal  in- 
cidence)   

45 

33 

19 

6 

-1 

-7 

_7 

2 

18 

POSSIBLE  LOWER  LIMITS  OF  SENSITIVITY.  On  certain  assumptions,  Sivian 
and  White  calculate  that  the  intensity  level  of  thermal-acoustic  noise,  i.e.,  noise  originat- 
ing from  the  thermal  velocities  of  air  molecules,  has  a  value  of  11  db  below  10  ~16  watt  for 
a  frequency  band  from  1000  to  6000  cycles,  and  hence  is  of  the  order  of  the  maximum  ear 
sensitivity.  Although  the  authors  regard  the  calculation  as  very  approximate,  it  suggests 
that  the  limit  of  sensitivity  may  be  set  by  the  transmitting  medium.  If  this  is  so,  man 
may  have  maximum  sensitivities  comparable  with  those  of  animals. 

EAR  SENSITIVITY  OF  THE  POPULATION.  The  values  of  ear  sensitivity  shown 
in  TaHe  1  are  for  a  selected  group  of  young  people  with  excellent  hearing;  not  all  people 


SENSITIVITY  OF  THE  EAR 


12-07 


hear  this  well.  During  the  New  York  and  San  Francisco  World  Fairs  in  1939,  which  were 
attended  by  people  from  all  parts  of  the  country,  a  survey  of  hearing  was  made  in  a  large 
group  of  the  United  States  population.  The  results,  which  are  plotted  in  Fig.  8,  show  the 
percentages  of  thresholds  measured  which  were  above  the  levels  shown  by  the  contour  lines, 
inus  tne  data  show  the  percentage  of  people  in  the  population  within  the  age  range 
irom  1U  to  59  years  who  cannot  hear  tones  below  the  levels  indicated  by  the  contours. 


10 


\ 


—ISO 


22,00-oj 


—120 


-60 


500^ 


300  r 


6O 


120 


180 


0 
Degrees 

FIG.  7a.    Increase  of  Monaural  Acuity  in  a  Free  Sound  Field  When  the  Direction  Which  the  Observer 

Faces  Is  Changed 


20 
15 

XO 

5 

1° 

O 
-5 

-ao 


—180 


7600 


10,OOC 


\5000x» 


\ 


\/ 


—120 


-60 


60 


120 


180 


0 
Degrees 

FIG.  76.     Increase  of  Monaural  Acuity  in  a  Free  Sound  Field  When  the  Direction  Which  the  Observer 

Faces  Is  Changed 

For  instance,  25  per  cent  of  the  population  cannot  hear  a  1000-cycle  tone  if  the  intensity 
is  below  the  20-db  level  (zero  =  10 ~16  watt  per  sq  cm).  The  dotted  portions  represent 
extrapolations  of  the  distributions  beyond  the  intensity  and  frequency  ranges  used  in  the 
tests  and  are,  of  course,  speculative. 

VARIATION  WITH  AGE.  Further  analysis  of  the  data  revealed  a  progressive  dete- 
rioration of  hearing  with  age  for  the  average  individual,  particularly  at  the  higher  fre- 
quencies. The  hearing  loss  is  somewhat  greater  for  men  than  for  women,  as  shown  by 
the  curves  in  Fig.  9.  For  this  plot  "zero  hearing  loss"  refers  to  an  ear  sensitivity  equal  to 
the  median  value  for  the  population,  tested.  The  contour  marked  "50  per  cent"  in  Fig.  8 
shows  the  median  values  as  a  function  of  frequency.  When  the  hearing  loss  exceeds  about 
25  db  at  all  frequencies  lower  than  2000  cycles,  a  person  will  experience  difficulty  at  times 
in  understanding  speech  in  classrooms,  auditoriums,  churches,  and  similar  environments 
where  the  speech  level  is  not  very  high.  A  hearing  loss  of  45  db  for  frequencies  below 
2000  cycles  results  in  some  difficulty  in  understanding  speech  at  distances  greater  than. 


12-08 


ACOUSTICS 


140 


?120 


100 


80 


60 


40 


.£      0 


\ 


—20 


\ 


\ 


Feeling 


Percent 
1 


20  50        100       200  500      1000     2000         5000    10,000  20,000 

Frequency  In  Cycles  per  Second 

FIG.  8.     Contour  Lines  Showing  the  Lower  Limit  of  the  Sensitivity  for  a  Given  Percentage  of  Tests 
in  the  Age  Group  10-59.    (Steinberg,  Montgomery,  and  Gardner.) 


30 


880  1760  3520 

Frequency  In  Cycles  per  Second 


7040 


FIG.  9.    Average  Hearing  Loss  for  Men  and  Women  in  Three  Age  Groups.    (Steinberg,  Montgomery 

and  Gardner.)  ' 


DIFFERENTIAL  SENSITIVITY 


12-09 


than  25  db  and  45  db 

(Steinberg,  Montgomery,  and  Gardner) 


2  or  3  ft,  and  considerable  benefit  is  usually  derived  from  the  use  of  a  hearing  aid.    Table 

3  shows  the  prevalence  of  hearing  losses  greater  than  25  and  45  db  among  the  people 
tested. 

In  the  age  group  from  50  to  59  years,  Table  3  shows  that  3  or  4  per  cent  of  the  popula- 
tion have  hearing  losses  exceeding  45  db  and,  therefore,  would  experience  some  difficulty 
understanding  speech  at  a  distance  of  2  or  3  ft.  Until  the  hearing  loss  exceeds  about  65  db, 
a  person  will  still  be  able 

to  use  the  telephone  with-   Table  3.     Percentage  of  Tests  with  Hearing  Losses  Greater 
out  much  difficulty. 

AUDITORY  RANGE. 
As  the  intensity  level  of  a 
sound  is  increased,  a  value 
is  reached  which  causes  a 
sensation  of  feeling  in  ad- 
dition to  the  sensation 
of  tone.  Such  intensity 
levels  may  be  taken  as 
practical  upper  limits  of 
hearing,  and  they  have 
been  called  the  "threshold 
of  feeling."  The  observed 
data  on  the  threshold  of 
feeling  were  the  number 
of  decibels  between  the 
hearing  and  feeling  thresh- 
olds for  pure  tones  and 
were  obtained  by  means 
of  telephone  receivers  held 
to  the  ear.  The  absolute 
threshold  of  f  eeling  plotted 
in  Fig.  8  was  obtained  by 
adding  the  observed  data  to  the  lower  limit  of  audibility  shown  in  Fig.  8.  The  lower 
limit  is  the  M.A.F.  determination  of  Table  1. 

The  lower  limit  and  the  feeling  curve  have  been  extrapolated  until  they  meet,  thus 
forming  an  enclosure  called  the  "auditory  sensation  area."  This  area  has  the  property 
that  any  sinusoidal  sound  wave  having  a  frequency  and  an  intensity  level  within  the  area 
will  cause  a  sensation  of  tone.  The  dashed  portions  of  the  curves  serve  as  a  means  of 
defining  the  upper  and  lower  frequency  limits,  i.e.,  the  lowest  and  the  highest  frequency 
that  can  be  sensed  as  a  tone.  Measurements  have  been  made  using  frequencies  varying 
from  about  8  to  40  cycles  for  the  lower  limit  and  from  12,000  to  35T000  cycles  for  the  upper 
limit,  but,  for  the  most  part,  very  little  attention  was  given  by  the  experimenters  to  the 
intensity  levels  at  which  the  determinations  were  made.  In  drawing  in  the  dashed  curves, 
consideration  was  given  to  available  data  on  frequency  limits,  and  values  of  20  cycles  and 
20,000  cycles  were  selected  for  the  lower  and  upper  frequency  limits,  respectively.  These 
limits  were  taken  as  more  or  less  typical  for  persons  of  normal  hearing  in  the  age  range 
from  18  to  23  years  but  do  not  represent  the  extreme  values  found  in  exceptional  cases. 


Age  Group 

25-dbLoss 

45-dbLoss 

Frequency 

Frequency 

880 

1760 

3520 

7040 

880 

1760 

3520 

10-19 

Men 

1.7 
1.8 

1.1 
1.8 

1.8 
3.5 

5.5 
7 

9.5 
13 

1.6 
1.2 

1.2 
1.6 

3.5 
3.5 

9.5 
7 

17 
14 

4.5 
1.2 

7 
2.2 

15 
5.5 

32 
11 

48 
22 

8 
2.5 

9.5 
3.5 

19 
10 

39 
24 

58 
43 

0.6 
0.6 

O.I 
0.4 

0.3 
1.2 

1.4 
2.1 

2.6 
4 

0.6 
0.4 

0.3 
0.3 

0.6 
0.8 

2.6 
1.5 

6 

3 

1.8 
0.3 

2.7 
0.7 

6 
1.6 

16 
3 

27 
7 

Women   . 

20-29 

Men 

Women   .  . 

30-39 

Men 

Women  

40-49 

Men 

Women       .... 

50-59 

Men 

Women  , 

3.  DIFFERENTIAL  SENSITIVITY 

The  differential  sensitivity  of  hearing  refers  to  some  aspect  of  the  smallest  change  of  a 
stimulus  that  can  be  detected.  Available  data  on  the  subject  are  confined  almost  entirely 
to  measurements  of  the  minimum  perceptible  increments  of  frequency  and  of  intensity 
when  the  stimulus  is  a  single-frequency  tone.  The  values  obtained  depend  to  some  extent 
upon  the  method  of  measurement,  probably  because  the  just  perceptible  change  (differ- 
ence limen)  is  affected  by  the  rate  at  which  the  change  is  made  as  well  as  other  variables 
common  to  tests  of  this  type.  Until  a  measuring  technique  is  standardized,  the  data  shown 
should  be  regarded  as  of  the  exploratory  type.  In  Fig.  10  the  results  of  extensive  measure- 
ments of  A/T  the  difference  limen  for  frequency,  are  plotted.  For  levels  greater  than  40  db 
above  threshold,  and  for  frequencies  greater  than  500  cycles,  the  frequency  D.L.  has  the 
approximately  constant  value  of  0.3  per  cent. 

These  results  were  obtained  by  listening  to  the  tones  with  head  receivers.  When  the 
measurements  are  made  by  listening  to  a  loudspeaker  in  a  room,  much  smaller  values 
can  be  observed.  In  this  case  the  change  in  frequency  is  detected  by  intensity  changes 
due  to  the  shifting  of  the  interference  pattern  in  the  room  with  frequency  change.  A 


12-10 


ACOUSTICS 


100 


1000 


10,000 


Frequency  In  Cycles 


FIG.  10.    The  Frequency  Difference  Limen  with  Decibels  above  Threshold  as  a  Parameter.    (Shower 

and  Biddulph.) 

similar  set  of  measurements  of  AI,  the  minimum  perceptible  increment  of  intensity,  is 
shown  in  Fig.  11.  For  convenient  use  the  increment  of  intensity  expressed  in  decibels, 
10  log  (I  +  AJ/I),  is  plotted  as  the  ordinate.  For  levels  greater  than  40  db  above  thresh- 
old, and  for  frequencies  between  200  and  7000  cycles,  the  intensity  D.L.  varies  from  0.25 
to  0.75  db. 


10.0 


O.I 


100 


1000 


10,000 


Frequency  in  Cycles 
It    The  Intensity  Difference  Limen  with  Decibels  above  Threshold  as  a  Parameter.    (Biesz.) 


LOUDNESS  OF  BOUNDS 


12-11 


4.  MASKING  EFFECTS  OF  SOUNDS 

When  listening  to  speech  or  music  it  often  occurs  that  a  disturbing  noise  interferes  to 
such  an  extent  that  the  desired  sounds  are  partially  or  entirely  obliterated.  The  noise  is 
said  to  have  a  masking  effect,  and  the  magnitude  of  the  effect  is  defined  by  a  "masking 
spectrum"  of  the  noise.  Measurements  of  masking  spectrums  are  made  by  determining  the 
threshold  of  audibility  of 
single-frequency  tones  in  "^  90 
the  presence  of  the  noise 
and  again  when  the  noise 
is  absent.  If  0  is  the  in- 
tensity level  of  a  tone  of 
frequency  /,  that  is  just 
audible  in  the  presence  of 
a  noise,  and  jSQ  is  the  in- 
tensity level  that  is  just 
audible  under  quiet  listen- 
ing conditions,  then  M, 
the  masking  at  the  fre- 
quency /,  is  denned  by  the 
equation :  M  —  fi  —  j8o.  A 
plot  of  M  as  a  function  of 
the  frequency  of  the  tones 
is  called  a  masking  spec- 
trum. If  the  intensity 
spectrum  of  a  noise  has 
been  measured  by  means 
of  a  sound  meter  and 
niters,  the  masking  spec- 
trum may  be  derived  by 
use  of  Fig.  12.  Here  the 
iso-masking  intensity  per 
cycle  level  of  noise  has 


=-30 


FIG.  12. 


100  500         1000 

Frequency  In  Cycles  per  Second 

Masking  Contours  for  a  Steady    Noise. 
Munson.) 


100OO 


(Fletcher     and 


been  plotted  as  a  function  of  frequency  with  masking  as  a  parameter.  The  applications  of 
these  data  are  limited  to  portions  of  the  noise  intensity  spectrum  that  do  not  exhibit  the 
abrupt  changes  with  frequency  that  occur  when  a  prominent  single-frequency  component 
is  present  or  a  filter  with  a  sharp  cutoff  limits  the  frequency  range  of  the  noise. 


5.  LOUDNESS  OF  SOUNDS 

Loudness  is  denned  as  the  magnitude  of  an  auditory  sensation,  and  for  steady  sounds 
it  is  thought  to  be  proportional  to  the  rate  at  which  neural  pulses  originating  along  the 
basilar  membrane  arrive  at  the  brain.  A  scale  of  auditory  magnitudes  has  been  derived 
from  loudness  tests  and  can  be  used  whenever  the  loudness  level  of  a  sound  is  known.  A 
measurement  of  loudness  level  consists  of  a  listening  test  in  which  the  level  of  a  1000- 
cycle  reference  tone  is  adjusted  until  it  sounds  equally  loud  to  the  sound  being  measured. 
Errors  in  judgment  may  be  large,  and  a  number  of  observers  may  be  required  to  obtain  a 
reliable  average  result.  The  equivalent  free  field  intensity  level  of  the  equally  loud  1000- 
cycle  tone  is,  by  definition,  the  loudness  level  of  the  unknown  sound. 

The  loudness  level  of  a  sound  being  known,  its  auditory  magnitude  is  found  by  referring 
to  Table  4,  which  gives  the  auditory  magnitude,  or  loudness,  as  a  function  of  the  intensity 
level  of  the  equally  loud  1000-cycle  tone.  The  figures  for  the  intensity  level  (loudness 
level)  of  the  1000-cycle  tone  are  in  decibels  relative  to  10~16  watt  per  sq  cm,  and  a  level  of 
40  db  results  in  an  auditory  magnitude  of  one  sone.  Thus,  in  Table  4,  the  loudness  cor- 
responding to  a  loudness  level  of  40  db  is  1  loudness  unit  (1  sone  =  1000  millisones).  In- 
creasing the  loudness  level  from  40  db  to  49  db  gives  a  listener  the  impression  that  the 
magnitude  has  doubled,  so  a  loudness  level  of  49  db  results  in  a  loudness  of  2  units.  In- 
creasing the  loudness  level  from  49  to  58.1  db  again  doubles  the  magnitude  of  the  sensation 
for  the  average  listener,  so  a  loudness  level  of  58.1  db  produces  a  loudness  of  4  units.  The 
empirical  relationship  between  loudness  and  loudness  level  is  shown  in  Table  4. 

Loudness  levels  are  given  in  steps  of  1  db,  or  1  "phon,"  and  zero  level  is  10  16  watt  per 
sq  cm  in  a  free  sound  field.  The  term  "phon"  is  generally  used  as  the  unit  of  loudness 
level  to  avoid  confusion  with  intensity  levels  of  sounds  other  than  a  1000-cycle  tone.  It 


12-12 


ACOUSTICS 


is  believed  that  the  loudness  of  a  steady  sound  depends  upon  the  rate  at  which  neural 
pulses  reach  the  brain,  and,  although  no  measurements  are  available  to  substantiate  this 
hypothesis,  the  concept  is  useful  in  explaining  empirical  methods  of  computing  the  loud- 
ness  of  complex  sounds.  It  follows  from  the  manner  in  which  the  loudness  scale  was  de- 
veloped that  a  1-sone  sound,  for  example,  is  equally  loud  to  any  other  1-sone^sound,  and 
four  times  as  loud  as  a  0.250-sone  sound.  It  is  also  true  that  a  1-sone  sound  will  drop  to  a 

Table  4.    Loudness  (Millisones)  vs.  Loudness  Level  (Pkons) 

(Fletcher  and  Munson) 
Loudness 


Loudness 
Level,  db 

0 

1 

2 

3 

4 

5 

6 

7 

8 

9 

0 

1.0 

1.42 

1.95 

2.58 

3.36 

4.32 

5.57 

7.10 

9.00 

11.4 

10 

14.4 

18.7 

23.3 

28.9 

35.1 

42,2 

50.6 

60.3 

71.6 

85.0 

20 

100 

120 

142 

165 

188 

214 

242 

272 

307 

340 

30 

380 

421 

470 

522 

577 

635 

700 

763 

835 

915 

40 

1,000 

1,080 

1,170 

1,260 

1,360 

1,470 

1,590 

1,710 

1,850 

2,000 

50 

2,150 

2,330 

2,510 

2,710 

2,930 

3,160 

3,410 

3,690 

3,980 

4,300 

60 

4,640 

5,010 

5,410 

5,840 

6,310 

6,810 

7,360 

7,940 

8,580 

9,260 

70 

10,000 

10,800 

11,700 

12,600 

13,600 

14,700 

15,900 

17,100 

18,500 

20,000 

80 

21,500 

23,300 

25,  IQO 

27,100 

29,300 

31,600 

34,100 

36,900 

39,800 

43,000 

90 

46,400 

50,100 

54,100 

58,400 

63,  100 

68,100 

73,600 

79,400 

85,800 

92,600 

100 

100,000 

108,000 

117,000 

126,000 

136,000 

147,000 

159,000 

171,000 

185,000 

200,000 

100 


80 


half  sone  if  only  one  ear  is  used  for  listening.  These  simple  relations  have  all  been  verified 
experimentally.  A  complex  sound  having  two  1-sone  components  will  be  equally  loud  to 
a  sound  of  2  sones  if  the  components  excite  nerve  endings  located  in  different  sections  of 
the  basilar  membrane  and  thus  contribute  1  sone  each,  even  though  they  are  sounded 
simultaneously.  This  is  not  likely  to  be  the  case  when  the  frequencies  of  the  components 
are  close  together  or  the  levels  are  high,  since  many  of  the  same  nerve  endings  are  then 
used  by  both  components. 

EFFECTIVE  STIMULATION  DENSITY.  The  response  of  the  auditory  nerve  endings 
is  dependent  upon  the  density  of  stimulation.  If  the  stimulus  is  localized,  the  response 
will  be  different,  and  the  loudness  sensation  will  differ,  in  general,  from  the  loudness 
resulting  when  the  stimulus  energy  is  distributed  among  a  large  number  of  nerve  endings. 

The  relationship  between 
the  frequency  of  the  stim- 
ulus and  the  position  co- 
ordinate (X)  of  the  nerve 
endings  stimulated  is 
shown  in  Fig.  13. 

The  ordinate  of  Pig.  13 
gives  the  position  of  max- 
imum stimulation  of  the 
nerve  endings,  with  re- 
spect to  the  total  number, 
when  the  stimulus  is  a 
single-frequency  tone. 
For  instance,  at  1000 
cycles  the  curve  shows 
that  31  per  cent  of  the 
nerve  endings  are  on  one 
side  of  the  point  of  max- 
imum stimulation,  and  69 
per  cent  are  on  the  other 


100 


10000    20000 


1000 
Frequency  In  Cycles  per  Second 

FIG.  13.    Relation  between  Frequency  of  the  Stimulus  and  the  Position 
of  Maximum  Stimulation.    (Fletcher.) 


side.  The  relationship  shown  can  be  derived  in  several  different  ways  on  the  basis  of  dif- 
ferent assumptions,  but  it  has  been  verified  only  in  a  qualitative  sense  by  actual  nerve  counts. 
To  obtain  the  effective  stimulation  density  as  a  function  of  frequency,  for  a  sound  with  a  con- 
tinuous energy  spectrum,  measurements  must  first  be  made  with  a  sound  meter  that  will 
analyze  sound  at  all  frequencies  within  the  audible  range.  If  the  analyzer , measures  the 
intensity  (I)  in  a  frequency  band  A/  cycles  wide,  then  the  readings  are  related  to  effective 
stimulation  densities  by  means  of  the  equation: 

Z  =  10  log 


LOTJDNESS  OF  SOUNDS 


12-13 


where  /i  is  the  intensity  level  of  a  1-millisone  single-frequency  tone,  and  A/i  is  a  frequency 
band  enclosing  a  unit  group  (1  per  cent)  of  the  nerve  endings.  It  is  seen  that  Z  is  the 
ratio,  in  decibels,  of  the  stimulation  intensity  per  unit  group  of  nerve  endings  to  the 
intensity  required  to  excite  a  1-miUisone  response  from  the  nerves.  The  equation  for  Z  is 
usually  given  in  the  more  convenient  form: 

Z  =  B  -f-  *  -  ft, 

where  B  is  the  measured  intensity  per  cycle  level  of  noise  [B  =  10  log  (J//0  A/)],  IQ  is  the 
reference  intensity  of  10~16  watt  per  sq  cm,  K  is  the  band  width  in  decibels  which  includes 
a  unit  nerve  group  (K  =  10  log  A/0,  and  ft,  is  the  intensity  level  of  a  tone  when  the  loud- 
ness  level  is  zero  [&>  =  10  log  (/i/I0)].  Values  of  K,  B0,  and  X  are  shown  in  Table  5  for  use 
in  computing  and  plotting  the  effective  stimulation  density  for  the  nerve  endings  as  a 
function  of  their  position  coordinate,  X. 

Table  5.    Values  of  K,  ft,  and  X 

(Fletcher  and  Munson) 

/          100        200        300        500        700          1,000        2,000        3,000  5,000  7,000  10,000 

X         1             4             7           14           21               31               52               64               78  85  92 

K         16.5       15.0       15.0       15.0      15.3          16.0          18.6          20.7          23.9  26.1  28.6 

j30      37.4      22.9       14.6        5.7        2.0            0.0         -4.5         -8.5  -4.9  4,9  8.8 

LOUDNESS    COMPUTATION    FOR    SOUNDS    WITH    CONTINUOUS    ENERGY 
SPECTRUMS.     The  loudness  of  a  c 

steady  sound  is  believed  to  be  propor- 
tional to  the  number  of  nerve  pulses 
arriving  at  the  brain  in  unit  time 
from  all  parts  of  the  basilar  mem- 
brane. We  may  compute  the  loud- 
ness  (AO  from  the  expression: 


N 


/•100 

*  f   * 

*/0 


NxdX 


where  NX>  the  sone  density,  is  a 
measure  of  the  number  of  nerve 
pulses  originating  from  a  unit  group 
of  nerve  endings  in  unit  time,  and 
X  is  the  position  coordinate  pre- 
viously denned.  The  equation  is 
solved  by  plotting  NX  as  a  function 
of  X  and  measuring  the  area  under 
the  curve.  Thus  the  loudness,  N1  of 
any  sound  can  be  computed  when  the 
sone  density  is  known  as  a  function 
of  X. 

The  relationship  between  sone 
density  (Nx)  and  the  effective  stim- 
ulation density  (Zx)  has  been  inves- 
tigated for  sounds  characterized  by  a 
continuous  distribution  of  energy, 
and  it  is  shown  in  Fig.  14. 

It  is  not  applicable  to  sounds  hav- 
ing single-frequency  components  or 
sounds  in  which  the  value  of  the 
slope,  dZ/dX,  exceeds  the  limits 
±2  db.  In  the  latter  case,  masking 
measurements  are  a  more  reliable 
means  of  obtaining  the  sone  density 
(A^x).  The  masking  (M)  is  defined 
by  the  equation :  M  —  $  —  ft,,  where 
£  is  the  intensity  level  of  a  single- 
frequency  tone  that  is  just  audible 
in  the  presence  of  a  masking  sound, 
and  /?o  is  the  intensity  level  when 
the  loudness  level  is  zero.  Figure 
15  shows  the  relationship  between 


« 


±'10 


.310" 


10 


jz: 


7 


-20  -10     0     10    20    30    40     50    60    70    80 

Zx.  Elective  Stimulation 
per  Unit  Nerve  Group  (db) 

FIG.  14.     Relation  between  the  Loudness  (Nx)  per  Unit 

Nerve  Group  and  the  Stimulation  (Zx)  per  Unit  Nerve 

Group.     (Fletcher  and  Munson.) 


12-14 


ACOUSTICS 


masking  and  sone  density.  The  masking  method  of  obtaining  the  sone  density  is  valid 
for  sounds  exhibiting  large  values  of  dZ/dX  but  not  for  sounds  with  single-frequency 
components.  The  masking  spectrum  of  a  single-frequency  tone  is  not  an  accurate  indica- 
tion of  its  sone  density  since  the  phenomenon  of  beats  between  the  masking  tone  and  the 
masked  tone  changes  the  conditions  under  which  the  masked  tone  is  detected.  This  re- 
sults in  low  values  of  masking  at  frequencies  where  the  highest  values  would  be  expected. 


10 


10 


10 


z 


0      10    20    30   40     50    60    70    80    90   100 
Masking,  M  (db) 

FIG.  15.    Relation  between  Loudness  Contribution  (JVx)  per  Unit  Nerve  Group  and  Masking  (Af). 

(Fletcher  and  Munson.) 

LOUDNESS  OF  SOUNDS  COMPOSED  OF  SINGLE-FREQUENCY  COMPO- 
NENTS. When  a  sound  is  a  single-frequency  tone  at  a  low  level,  the  nerve  endings  in 
a  localized  region  of  the  basilar  membrane  are  stimulated.  As  the  level  is  increased,  the 
stimulation  pattern  spreads  to  adjacent  nerves,  and  at  high  levels  a  large  proportion  of 
all  the  nerves  may  come  into  use.  The  region  of  maximum  stimulation  probably  does  not 
change  much  when  the  level  is  raised,  but  secondary  maxima  appear  at  harmonic  fre- 
quencies owing  to  non-linear  distortion  in  the  ear  itself.  The  masking  spectrums  shown  in 
Fig.  16  indicate  the  nature  of  the  change  of  the  stimulation  pattern  as  the  level  of  a  tone 
is  increased. 

As  explained  in  the  previous  section,  the  regions  of  maximum  stimulation  appear  as 
depressions  because  of  the  beats  which  occur  near  the  fundamental  and  harmonic  fre- 
quencies. Measurements  of  the  conditions  for  best  beats  have  been  used  to  determine  the 
magnitude  of  harmonics  produced  in  the  ear.  Figure  17  shows  the  equivalent  magnitude  of 
the  subjective  harmonics  for  different  pressure  levels  of  the  applied  sound.  Taking  the 
fundamental  as  the  first  harmonic,  the  abscissa  gives  the  number  of  the  harmonic  When 


LOTJDNESS  OF  SOUNDS 


12-15 


plotted  in  terms  of  pressure  level,  the  curves  are  independent  of  frequency.    It  is  clear 
that  the  stimulation  patterns  of  single-frequency  tones  are  very  complex,  and  it  would  be 
difficult  to  plot  the  sone  density  as  a  function  of  X,  as  was  done  for  sounds  with  continu- 
ous energy  spectrums.   However,  the  loud- 
ness   of   any  single  component 
found  from  the   loudness-level 
shown  in  Fig.  18. 


may   be 
contours 


100 
90 


70 


800  Cycles 


\\.s 


80 


8888 

00        W        ID         O 

w      to      co      •* 
Frequency  tn  Cycles  per  Second 

FIG.  16.     Masking  Spectrums  of    a  Single-fre- 
quency Tone  with  Level  above  Threshold  as  a 
Parameter.    (Fletcher,  Wegel,  and  Lane.) 


g  140 

tr 
0  130 

JS 
i 

\! 

! 

c 

0. 

o  1  2n 

t* 

\! 

N 

V    :         1 

i 

1 

T3 

i" 

\r 

\i 

X 

o  110 

8. 

f 

\> 

\ 

X  ; 

N 

XV 

.Q 

"°    on 

[ 

\> 

—  x 

\i  1 

\ 

^^ 

» 

o    9O 
I 

c      or. 

f 

O 

s^ 

\l\ 

X 

I  80 

-a 

} 

\\ 

\ 

k\ 

\ 

s 

1 

r- 

- 

5     70 
o 
•=     ^« 

\\ 

\ 

\  s 

K 

^s 

N 

-    6O 

\\ 

\ 

\\ 

K 

1 

1  M 

g     .f. 

\ 

s1 

KK 

!\ 

v 

l\ 

D     40 

JE 

\ 

,\ 

1   Sl    " 

1 
\ 

1^ 

4, 

N 

^ 

0       3O 

\ 

\  \ 

\ 

K 

\ 

N 

\ 

o    2° 

1     ,A 

\ 

*  \ 

\ 

\ 

>, 

< 

Number  of  Harmonic 

FIG.    17.    Magnitude  of  Subjective  Harmonics. 
(Fletcher  and  Graham.) 


The  ordinate  here  is  the  free  field  intensity  level  of  a  pure  tone,  and  each  contour  line 
is  marked  with  a  loudness  level.  For  instance,  a  200-cyele  tone  at  an  intensity  level  of 
60  db  from  ICT16  watt  has  a  loudness  level  of  50  db.  Turning  to  Table  4,  we  see  that  the 
corresponding  loudness  is  2.150  sones.  If  a  sound  has  more  than  one  component,  the  sone 
values  may  be  added  to  obtain  the  total  loudness,  provided  that  the  frequencies  of  tne 


120 


rtfl  100  =MJV  iWU 

Frequency  in  Cycles  per  Second 
FIG.  18.    Loudness-level  Contours.    (Fletcher  and  Munson.) 

components  differ  enough  so  that  different  nerve  groups  are  stimulated.    The  , 
r^quLd  is  a  complex  function  of  level  and  frequency  and  will  not  be  discussed 
Some  idea  of  the  effect  of  different  separations  of  components  may  be  obtained  from 


12-16 


ACOUSTICS 


Fig.  19,  which  shows  the  loudness  levels  of  several  10-eomponent  sounds  as  a  function  of 
the  loudness  level  of  each  component. 

The  first  sound  had  a  fundamental  frequency  of  530  cycles  and  a  difference  oi  5^0 
cycles  between  components.  The  others  had  a  fundamental  of  1000  cycles  and  the  dif- 
ferences indicated.  The  dotted  line  shows  the  loudness  levels  corresponding  to  10  times 
the  loudness  of  a  single  component.  For  a  more  extended  treatment  of  the  loudness  of 
sounds  with  single-frequency  components,  refer  to  an  article  in  the  October  1933  Journal 
of  the  Acoustical  Society  of  America  entitled  "Loudness,  Its  Definition,  Measurement  and 
Calculation,"  by  Fletcher  and  Munson. 


Loudness  Level  of  10  Components,  db  ^ 

Mtow^tncn^ooioo 
ooo  Ooo  ooooo 

s// 

^/ 

X 

/& 

% 

/ 

//, 

Y/ 

/ 

& 

Y' 

/* 

t 

V/ 

/ 

I 

/ 

A 

/ 

4 

'A 

a  Frequency  Difference  —  530 
o         "                 "         =340 
A         "                 "         =230 
x         ,<                 "         =112 
.         <.                 "         =50 

/ 

•/ 

-10° 


80        90 


10        20        30        40        50        60         70 
Loudness  Level  of  Single  Component,  db 

FIG.  19.     Effect  of  Separation  of  Components  on  the  Loudness  Levels  of  Complex  Tones.    (Fletcher 

and  Munson.) 


6.  THE  PITCH  OF  STEADY  SOUNDS 

The  pitch  of  a  steady  sound  is  the  position  on  a  musical  scale  that  would  be  assigned  to 
it  by  a  listener.  If  the  sound  is  a  single-frequency  tone,  its  pitch  depends  upon  frequency 
and,  to  a  slight  degree,  upon  intensity.  At  loudness  levels  less  than  40  db  the  pitch  de- 
pends only  upon  frequency,  and  single-frequency  tones  at  a  loudness  level  of  40  db  have 
been  chosen  as  standards  for  comparison  with  sounds  of  unknown  pitch.  The  results  of 
pitch  comparisons  between  single-frequency  tones  at  a  40-db  loudness  level  and  tones  at 
higher  levels  are  shown  in  Fig.  20.  The  ordinate  is  the  change  in  frequency,  in  per  cent, 
of  a  tone  that  is  necessary  in  order  that  its  pitch  remain  constant  as  its  level  is  raised. 

For  example,  the  curves  show  that  a  100-cycle  tone  must  be  lowered  10  per  cent  when 
the  loudness  level  is  increased  from  the  standard  level  to  100  db.  This  means  that  a 
90-cycle  tone  at  a  loudness  level  of  100  will  appear  to  have  the  same  pitch  as  a  100-cycle 
tone  at  a  loudness  level  of  40  db.  The  curves  shown  in  Fig.  20  are  based  on  data  at  low 
frequencies.  Similar  experiments  by  Zurmuhl  and  Stevens  indicate  that  at  frequencies 
above  2000  cycles  there  is  a  small  increase  in  pitch  as  the  level  is  raised.  An  observer's 
judgment  of  the  pitch  of  a  sound  is  thought  to  be  related  to  the  position  of  the  stimulated 
region  on  the  basilar  membrane.  In  general,  a  sensation  of  lower  pitch  occurs  when  the 
stimulated  region  shifts  towards  the  helicotrema. 

Table  6.    Judgments  of  Half  Pitch 
(Stevens  and  Volkmann) 

Standard  frequency 150       250       500       1,000       2,000       3,000       5,000       10,000 

Frequency  far  half  pitch 85       111       206          373  633       1,009       1,437         2,064 


THE  PITCH  OF  STEADY  SOUNDS 


12-17 


Experiments  have  been  made  to  determine  how  much  the  frequency  of  a  standard  tone 
must  be  shifted  to  result  in  the  sensation  that  the  pitch  has  decreased  by  a  factor  of  one- 
half.  The  results  are  shown  in  Table  6,  where  the  first  row  is  the  frequency  of  the  standard 
tone.  The  second  row  is  the  mean  frequency  of  tones  which  were  selected  by  twelve 


-25 


100 


ZOOG 


200  5OO 

Ereque'ncy  In  Cycles  per  Second 

FIG.  20.     Pitch  Change  of  Single-frequency  Tones  at  High  Levels.    (Snow.) 


2000 


observers  to  be  one-half  the  pitch  of  the  standard.  The  loudness  level  of  all  tones  was 
40  db.  Table  6  shows  that  a  tone  of  373  cycles  will  appear  to  be  one-half  as  high  in  pitch 
as  1000  cycles.  By  use  of  these  and  other  data  on  estimations  of  pitch  intervals,  Stevens, 
Volkmann,  and  Newman  have  devised  a  numerical  pitch  scale  having  the  property  that 
tones  which  appear  to  be  50  per  cent  lower  in  pitch  will  also  be  related  in  the  same  manner 
on  the  pitch  scale.  The  unit  of  pitch  is  called  a  *'melT"  and  the  relationship  between  mels 
and  frequency  for  pure  tones  at  a  loudness  level  of  40  db  is  shown  in  Fig.  21. 


BOOO 


20 


100       200       400  1000    2000      4OOO       10,OOO 

Erequency 

FIG.  21.     Relation  of  the  Pitch  in  Mels  to  the  Frequency  of  Tones  at  a  Loudness  Level  of  40  Db. 

(Stevens  and  Volkmann.) 


12-18  ACOUSTICS 


7.  LOCALIZATION  OP  SOUNDS 

The  ability  to  localize  the  direction  and  to  form  a  judgment  of  the  distance  away  of  a 
source  is  a  matter  of  common  experience,  but  very  few  quantitative  data  on  the  subject 
are  available.  The  localization  of  direction  (angular  localization)  appears  to  depend  upon 
the  detection  of  phase  differences  at  the  two  ears,  loudness  differences,  differences  in 
quality,  and  differences  in  arrival  times  at  the  two  ears.  Loudness  differences,  and  quality 
differences  (in  complex  sounds) ,  and  to  some  extent  phase  differences,  arise  from  diffrac- 
tion effects  of  the  head.  At  low  frequencies  the  loudness  difference  at  the  two  ears  is 
small r  but  localization  by  phase  difference  is  effective.  An  angular  accuracy  of  about 
±10°  is  obtained  for  sounds  directly  in  front  or  in  back  of  the  observer.  Much  less  preci- 
sion is  obtained  when  the  source  is  at  one  side.  Loudness  differences  are  most  effective 
in  the  frequency  range  above  3000  cycles  and  quality  differences  in  the  range  above  1000 
cycles.  In  general,  complex  sounds  having  prominent  high-frequency  components,  hence 
large  loudness  and  quality  differences  between  the  ears,  are  localized  with  greatest  accu- 
racy. Under  familiar  acoustic  conditions,  such  sounds  may  be  localized  frequently  by 
the  use  of  one  ear  only.  Presumably  this  is  accomplished  by  recognizing  the  characteristic 
distortion  introduced  by  the  head.  Steinberg  and  Snow  report  that  the  apparent  distance 
of  the  sound  source,  i.e.,  depth  localization,  depends  upon  the  loudness  of  a  sound.  When 
an  observer  is  listening  in  a  room,  the  ratio  of  the  direct  sound  intensity  (that  reaching 
the  ears  without  reflection)  to  the  reflected  sound  intensity  also  enters  into  depth  localiza- 
tion. 

BIBLIOGRAPHY 

American  Standards  Association,  American  Standard  Acoustical  Terminology,  J.  Acous.  Soc.  Am.f 

Vol.  14,  84  (July  1942). 
Beasley,  W.  C.,  Characteristics  and  Distribution  of  Impaired  Hearing  in  the  Population  of  the  U.  S., 

J.  Acous.  Soc.  Am.,  Vol.  12,  114  (July  1940). 

Bekesy,  G.,  Clicks  and  the  Theory  of  Hearing,  Physik.  Z&itschrift,  August  1934,  p.  577. 
Dimmick  and  Olson,  Intensive  Difference  Limen  in  Audition,  J.  Acous.  Soc.  Am.,  Vol.  12,  517  (April 

1941). 
Fletcher,  H.,  Loudness,  Masking  and  Their  Relation  to  the  Hearing  Process  and  the  Problem  of  Noise 

Measurement,  /.  Acous.  Soc.  Am.,  Vol.  9,  275  (April  1938). 
Fletcher,  H.,  Loudness,  Pitch,  and  Timbre  of  Musical  Tones,  J.  Acous.  Soc.  Am.,  Vol.  6,  59  (October 

1934). 

Fletcher,  H.T  A  Space-time  Pattern  Theory  of  Hearing,  J.  Acous.  Soc.  Am.,  April  1930,  p.  311. 
Fletcher,  H.,  Auditory  Patterns,  Rev.  Modern  Phys.,  Vol.  12,  47  (January  1940). 
Fletcher,  H.,  Speech  and  Hearing.     Van  Nostrand  (1929). 

Fletcher,  H.,  The  Mechanism  of  Hearing,  Proc.  Nail.  Acad.  Sci.,  Vol.  24,  265  (July  1938). 
Fletcher  and  Munson,  Loudness,  Its  Definition,  Measurement,  and  Calculation,  J.  Acous.  Soc.  Am.* 

Vol.  5,  82  (October  1933). 
Fletcher  and  Munson,  Relation  between  Loudness  and  Masking,  J.  Acous.  Soc.  Am.,  Vol.  9,  1  (July 

1937). 
Galambos  and  Davis,  The  Response  of  Single  Auditory-nerve  Fibers  to  Acoustic  Stimulation,  J.  Neuro- 

-physiol,  Vol.  6T  39  (1943), 
Inglis,  Gray,  and  Jenkins,  A  Voice  and  Ear  for  Telephone  Measurements,  Bell  Sys.  Tech.  J.,  April 

1932,  p.  293. 
Knudsen,  V.  O.,  Sensibility  of  the  Ear  to  Small  Differences  of  Intensity  and  Frequency,  Phys.  Rev., 

January  1923,  p.  84. 

Rawdon-Smith  and  Sturdy,  The  Effect  of  Adaptation  on  the  Differential  Threshold  for  Sound  In- 
tensity, Brit.  J.  Psych.,  Vol.  30,  124  (1939). 

Riesz,  R.  R.,  Differential  Intensity  Sensitivity'of  the  Ear  for  Pure  Tones,  Phys.  Rev.,  May  1928,  p.  867. 
Shower  and  Biddulph,  Differential  Pitch  Sensitivity  of  the  Ear,  J.  Acous.  Soc.  Am.,  October  1931, 

p.  275. 

Sivian  and  White,  On  Minimum  Audible  Sound  Fields,  J.  Acous.  Soc.  Am.,  April  1933,  p.  288. 
Snow,  W.  B.,  Change  of  Pitch  with  Loudness  at  Low  Frequencies,  J".  Acous.  Soc.  Am.,  Vol.  8,  14  (July 

1936). 
Steinberg,  J.  C.,  Positions  of  Stimulation  in  the  Cochlea  by  Pure  Tones,  /,  Acous.  Soc.  Am.,  VoL  8. 

176  (January  1937). 
Steinberg,  Montgomery,  and  Gardner,  Results  of  the  World's  Fair  Hearing  Tests,  J.  Acous.  Soc  Am 

Vol.  12,  291  (October  1940). 

Steinberg  and  Snow,  Auditory  Perspective — Physical  Factors,  Elec.  Eng.,  Vol.  53,  12  (January  1934). 
Stevens,  S.  S.,  The  Attributes  of  Tones,  Proc.  Natt.  Acad.  Sci.,  July  1934,  p.  457. 
Stevens,  S.  S.t  and  H.  Davis,  Hearing,  Its  Psychology  and  Physiology.    John  Wiley  (1938). 
Stevens  and  Davis,  Psychophysiological  Acoustics — Pitch  and  Loudness,  J.  Acous.  Soc.  Am  ,  Vol   8 

1  (July  1936). 
Stevens  and  Volkmann,  The  Relation  of  Pitch  to  Frequency:  A  Revised  Scale,  Am.  J.  Psych    Vol  53 

329  (July  1940).  '      ' 

Stevens,  Volkmann,  and  Newman,  Scale  for  the  Measurement  of  Pitch,  J".  Acous.  Soc.  Am.,  Vol.  8,  185 

(January  1937). 

Stewart,  G.  W.,  Localisation  of  Pure  Tones,  Phys.  Rev.,  May,  1920,  p.  425. 
Trimmer  and  Firestone,  An  Investigation  of  Subjective  Tones,  J.  Acous.  Soc.  Am.,  Vol   9   24  (July 

1937), 

Troger,  J.,  The  Reception  of  Sound  by  the  Outer  Ear,  Physik.  Zeitschrift,  January  1930,  p  26. 
Waliaeh,  Fam,  On  Sound  Localization,  /.  Acous.  Soc.  Am.t  Vol.  10,  270  (April  1939). 


PRODUCTION  OF  SPEECH  12-19 

Wegel  and  Lane,  Auditory  Masking  and  Dynamics  of  the  Inner  Ear,  Phys.  Rev.,  February  1924,  p.  266. 
Wever   E.G.,  The  Electrical  Responses  of  the  Ear,  Psych.  Bull.,  March  1939,  p.  36. 
Wnghton,  Sir  Thomas,  Analytical  Mechanism  of  the  Internal  Ear.     Macmillan,  London  (1918) 
Zurnmhl,  G.,  Variation  of  Pitch  Sensation  with  Loudness,  Zeit  f&r  Sinnesphysiologie,  August  1940, 

SPEECH  AND  MUSIC 

By  John  C.  Steinberg  and  W.  A.  Hanson 

8.  DESCRIPTION  OF  SPEECH  ORGANS 

The  speech  organs  consist  of  the  lungs  and  respiratory  muscles,  the  trachea  or  wind- 
pipe, the  larynx,  and  the  cavities  of  the  throat,  mouth,  and  nose.  In  speaking,  a  Sow  of 
air  is  produced  by  the  lungs  which  is  modified  in  passing  through  these  passages  to  form 
speech  sounds.  The  larynx  is  a  cavity  of  irregular  shape  formed  of  cartilage  located  at 
the  upper  end  of  the  trachea,  a  tube  some  12  cm  long  and  2  cm  in  diameter  leading  from 
the  lungs.  A  pair  of  muscular  ledges,  called  the  vocal  cords,  form  a  slit  in  the  larynx 
through  which  the  air  must  pass.  When  the  vocal  cords  are  set  in  vibration,  the  air  flow 
is  periodically  interrupted  and  the  sound  is  said  to  be  voiced.  Measurements  reported 
by  Riesz,  on  persons  whose  larynx  had  been  removed  by  a  surgical  operation,  indicate 
that  the  excess  pressure  in  the  lungs  when  producing  sustained  parts  of  speech,  such  as 
vowels,  is  of  the  order  of  4  mm  of  mercury  or  0.005  atmosphere.  The  rate  of  air  flow  is 
about  150  cu  cm  per  sec.  The  normal  capacity  of  the  lungs  is  about  2500  cu  cmT  and  the 
average  expiration  in  breathing  is  about  500  cu  cm. 

By  means  of  high-speed  motion-picture  photography,  views  of  the  vocal-cord  movements 
in  slow  motion  have  been  obtained  at  Bell  Telephone  Laboratories.  Measurements  on 
successive  exposures  of  the  vibrating  cords  indicated  a  displacement  amplitude  that  was 
sawtooth  in  form.  The  cords  tended  to  snap  apart  and  close  slowly  and  firmly  together. 
The  wave  form  contained  a  fundamental  and  higher  harmonics  which  diminished  in 
amplitude  inversely  as  the  square  of  the  harmonic  number.  The  length  of  the  cords  and 
maximum  amplitude  of  opening  varied  with  pitch.  For  example,  in  the  subjects  studied, 
the  cords  were  about  1.2  cm  in  length  and  their  widest  opening  was  about  0.4  cm  when 
vibrating  at  120  cycles.  At  300  cycles,  their  length  was  about  2.0  cm  and  the  widest 
opening  about  0.2  cm.  Changes  in  the  acoustic  load  by  closing  the  mouth  opening  with  a 
glass  window  or  filling  the  vocal  cavities  with  helium  did  not  appear  to  affect  the  vibration 
wave  form  of  the  vocal  cords  markedly. 

The  opening  between  the  cords  is  called  the  glottis,  which,  of  course,  varies  in  size  as 
the  cords  vibrate.  There  appears  to  be  little  direct  information  on  the  acoustic  wave  form 
produced  at  the  glottis  by  the  vocal-cord  vibrations.  Indirect  evidence  based  on  harmonic 
analysis  of  vowel  sounds  reported  by  Steinberg  and  also  Lewis  indicates  a  sawtooth  wave 
form  in  which  the  amplitudes  of  the  harmonics  dinunish  inversely  as  the  1,5  power  of  the 
harmonic  number. 

9.  PRODUCTION  OF  SPEECH 

MECHANISM  OF  SPEECH.  As  has  been  discussed  by  Dudley,  speech  may  be  re- 
garded as  a  phenomenon  of  modulation  in  which  the  breath  stream,  a  unidirectional  Sow 
of  air  produced  by  the  lungs,  is  given  intelligence-carrying  variations.  The  breath  stream 
is  made  audible  by  two  kinds  of  modulation  called  weal-cord  and  frictional  modulation. 
The  first  is  produced  by  the  periodic  interruption  of  the  breath  stream  by  the  vibrations 
of  the  vocal  cords.  The  second  is  produced  by  the  turbulent  flow  of  the  breath  stream 
through  constrictions  formed  in  the  vocal  tract.  Both  produce  variations  at  relatively 
high  rates,  i.e.,  in  excess  of  70  cycles.  The  vocal-cord  modulation  produces  fundamental 
and  harmonic  overtones  such  as  characterize  a  vocalised:  or  voiced  sound.  Frictional 
modulation  produces  a  wide  range  of  inharmonically  related  overtones  such  as  character- 
ize a  hiss. 

The  audible  components  thus  produced  are  in  turn  modulated  by  relatively  slow  varia- 
tions, i.e.,  at  rates  definable  by  frequency  components  in  the  range  below  some  40  or  50 
cycles.  One  of  these  is  called  start-stop  modulation,  which  is  accomplished  by  the  muscles 
of  respiration,  the  vocal  cords,  tongue,  and  lips.  Another,  called  cavity  modulation,  is 
produced  by  changes  in  size  and  shape  of  the  vocal  passages  extending  from  the  glottis  to 
the  mouth  and  nose  openings.  These  passages  selectively  transmit  and  radiate  the  audible 
frequency  components  produced  in  the  breath  stream  by  vocal-cord  and  frictional  rnodula- 


12-20 


ACOUSTICS 


tion.  This  results  in  a  reinforcement,  relatively,  of  certain  frequency  components  or 
regions  which  are  called  vocal  resonances. 

Several  other  types  of  low-frequency  modulation  occur  such  as  inflection,  vibrato,  and 
stress,  but  the  four  noted  above  are  the  ones  of  chief  importance  to  the  intelligibility  of 
American  speech  sounds.  Start-stop  modulation  is  one  of  the  principal  characteristics  of 
the  plosive  or  stop  consonant  group,  and  frictional  modulation  is  one  of  the  chief  charac- 
teristics of  the  fricative  consonant  group.  In  the  vowel  and  vowel-like  group  of  sounds, 
cavity  and  vocal-cord  modulation  are  among  the  conspicuous  characteristics. 

CHARACTERISTICS  OF  SPEECH  SOUNDS.  As  a  result  of  cavity  modulation,  all 
sounds  tend  to  show  characteristic  frequency  regions  of  reinforcement  in  a  greater  or  less 
degree.  These  vocal  resonances  are  more  conspicuous  in  the  vowel  and  vowel-like  sounds. 
In  general,  there  is  a  resonance  in  the  range  from  300  to  1000  cycles,  one  hi  the  range  from 
1000  to  2500  cycles,  and  one  or  more  in  the  range  from  2500  to  4000  cycles.  The  two 
resonances  below  2500  cycles  are  the  ones  that  vary  most  in  frequency  position  from 
sound  to  sound.  There  has  been  a  tendency  to  associate  the  low  resonance  with  the 
pharynx  or  throat  cavity  and  the  next  resonance  (1000-2500  cycles)  with  the  mouth 
cavity.  Damping  constants  reported  for  the  cavities  range  from  1000  to  4000  db  per  sec. 
With  a  vocal-cord  vibration  rate  of  125  cycles,  the  cavities  would  be  excited  periodically 
at  intervals  of  0.008  sec,  and  the  amplitude  of  the  sound  wave  would  be  expected  to  decay 
to  something  between  0.4  and  0.03  of  its  initial  value  during  the  interval. 

Table  1.    Approximate  Characteristics  of  Common  Speech  Sounds 


Sound 

fi 

Li 

h 

L2 

h 

Ls 

/4 

U 

n 

Pure  Vowels 
e  (eve) 

350 

ij 

2400 

-17 

3200 

-13 

3700 

-73 

6. 

44 

i  (it)        

450 

_4 

2150 

7950 

3600 

10. 

27 

e  (bet)  

550 

-3 

1950 

-9 

2700 

-13 

3700 

-19 

6. 

60 

a"  (at) 

800 

—  1 

1800 

2800 

3900 

6. 

89 

ah  (father)     .  . 

800 

0 

1250 

-8 

2950 

-22 

3800 

-30 

6. 

52 

a  fall) 

550 

0 

850 

2900 

4 

15 

o  (obey) 

450 

-2 

800 

-9 

2600 

-24 

3200 

—24 

4 

74 

u  (foot) 

400 

-2 

1050 

-12 

2300 

-24 

3200 

—34 

2 

96 

u  (boot)  

350 

—4 

950 

—19 

2250 

-32 

3200 

-34 

6. 

26 

Vowel-lite  Sounds 
1  (let) 

450 

—8 

1000 

—21 

2550 

-20 

2950 

—24 

tti 
4  31 

n/ 
8  40 

r  (run)  

500 

-5 

1350 

—12 

1850 

-16 

3500 

—29 

2  78 

13  05 

m  (me)  

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-11 

1250 

-21 

2250 

-23 

2750 

-30 

5.89 

5  48 

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u 

—  13 

1450 

-26 

2300 

-28 

2750 

—33 

4  99 

12  52 

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u 

-to 

2350 

2750 

3  57 

Fricative  Consonants 
v  (voice) 

Fundamental 

—  18 

1150 

-37 

2500 

3650 

12    (3  9} 

42    n  4} 

th  (that)        

—  18 

1450 

-27 

2550 

—45 

67    (20) 

I  2    (0  04) 

z  (zoo) 

u 

—  16 

2000 

—33 

2700 

—42 

03    (5  4^ 

6/1    c\  \\ 

zh  (pleasure)  

« 

-15 

2150 

—24 

2650 

-39 

0  02  (1  7) 

0  01  (0  3) 

Stop  Consonants 
b  (be)  

Timctamftntal 

—20 

800 

1350 

46    (25) 

04     (\    7\ 

d  (day)  

-20 

1700 

2450 

62    (78) 

4  4    M4^ 

g(get)  

fa 

17 

Variable 

> 

4.3    (5.5) 

0.4    (2.8) 

The  steady-state  resonance  characteristics  of  the  sounds  can  be  expressed  by  the  peak 
frequencies /t-  of  the  vocal  resonances,  the  levels  Li  of  the  peak  frequencies,  and  the  damp- 
ing constants  A*  expressed  in  decibels  per  second  which  indicate  the  sharpness  of  the 
resonances.  This  information,  together  with  the  data  on  the  acoustic  wave  form  at  the 
glottis  given  in  article  8,  permits  an  approximate  construction  of  the  spectrums  of  the  sounds. 
The  available  data  on  these  characteristics  are  summarized  in  Table  1.  The  columns 
designated  /i  to  /4  indicate  the  peak  frequencies  of  vocal  resonance.  They  were  obtained 
from  data  reported  by  Kopp  and  Green,  and  represent  values  for  one  voice.  Since  the 
values  vary  somewhat  from  voice  to  voice,  values  for  a  single  voice  were  chosen  to  permit 
a  relative  comparison  of  the  different  sounds.  When  "fundamental"  appears  in  the  col- 
umn fi,  it  means  that  the  resonance  was  too  close  to  the  fundamental  frequency  to  be 
resolved  by  the  analyzing  means.  The  vocal  resonances  shown  for  the  voiced  consonants, 
77,  th  (that),  z,  zh,  6,  d,  and  g,  apply  also  to  their  unvoiced  cognates,  /,  th  (thin),  s,  sh,  p, 
t,  and  k.  For  g  (get)  and  also  k  (key) ,  the  frequencies  of  the  vocal  resonances  vary  depend- 
ing upon  the  sound  with  which  they  are  combined.  The  column  LI  shows  the  relative 
levels  of  the  first  resonance  peaks  as  derived  from  Table  X,  Fletcher,  Speech  and 
Hearing. 


PKODUCTION  OF  SPEECH  12-21 

The  remaining  columns,  L2  to  L4f  show  levels  of  the  respective  resonant  peaks  based  on 
data  from  several  sources.  All  levels  are  expressed  in  decibels  below  the  level  of  the  first 
resonant  peak  in  ah  (father).  The  damping  constants  A,-  are  not  shown  but  vary  from 
about  1200  db  per  sec  for  the  first  resonance  to  about  4000  db  per  sec  for  the  fourth  res- 
onance. The  band  width  of  the  resonance  at  points  on  the  resonance  curve  3  db  below 
the  peak  response  is  given  approximately  by  A,-/27.3.  The  last  column,  designated  n, 
shows  the  relative  occurrence  of  the  sounds  in  telephone  conversation  from  a  report  by 
French,  Carter,  and  Koenig.  The  figures  indicate  the  percentage  occurrence  of  the  vowels 
shown  among  all  vowels  occurring  in  95,522  vowels.  The  occurrence  figures  are  indicated 
separately  for  initial  and  final  consonants  from  64,043  initial  and  65,544  final  consonants. 
The  figures  in  parentheses  indicate  the  occurrence  of  the  unvoiced  cognates  of  the  voiced 
consonants  shown.  The  figures  were  obtained  from  the  words,  syllables,  and  sounds 
occurring  in  telephone  conversations  after  the  exclusion  of  articles,  names,  titles,  exclama- 
tions, letters,  and  numbers. 

An  interesting  feature  of  the  report  is  the  extent  to  which  a  few  common  words  or 
sounds,  by  being  used  over  and  over,  from  a  large  part  of  ordinary  speech.  For  example, 
eight  different  words  and  four  different  sounds  account,  respectively,  for  25  per  cent  of 
the  total  words  and  sounds  used.  Thirty  words  and  ten  sounds  account,  respectively,  for 
50  per  cent  of  the  total  used. 

ARTIFICIAL  LARYNX.  The  artificial  larynx  is  based  on  the  principle  that  the  vocal- 
cord  tone  need  not  necessarily  arise  in  the  larynx  in  order  that  cavity  modulation  occur  to 
form  speech.  It  may  be  introduced  into  the  vocal  cavities  through  the  mouth  opening. 
A  means  for  accomplishing  this  has  been  described  by  Riesz  and  is  used  by  persons  who 
have  had  their  larynx  removed  by  surgery.  In  this  case,  the  windpipe  is  terminated  by 
a  small  opening  in  the  neck,  through  which  the  patient  breathes.  In  using  the  artificial 
larynx,  a  flexible  rubber  tube  is  fitted  over  the  neck  opening  and  leads  to  a  reed  which 
vibrates  with  the  passage  of  air.  The  vibrated  air  stream  is  led  to  the  mouth  opening  by 
a  flexible  tube,  and,  with  practice,  the  patient  learns  to  modulate  the  sound  and  form 
speech  by  the  ordinary  articulatory  movements.  Firestone  has  described  and  demon- 
strated a  form  of  artificial  larynx  for  introducing  various  types  of  sounds  through  the 
mouth  opening  for  vocal  modulation.  Mr.  G.  M.  Wright  has  developed  an  artificial 
larynx  in  which  sound  is  introduced  into  the  vocal  tract  by  means  of  vibrators  attached 
outside  the  throat.  The  device  has  been  used  to  produce  unusual  vocalised  sound  effects 
in  radio  work. 

ARTIFICIAL  VOICE.  An  artificial  voice  described  by  Inglis,  Gray,  and  Jenkins, 
designed  for  the  measurement  of  microphones,  consists  of  a  small  moving-coil  loudspeaker 
having  an  opening  somewhat  larger  than  that  of  the  human  mouth.  Its  design  provides 
for  an  undistorted  acoustic  power  output  comparable  with  powers  produced  in  speaking. 
Its  sound  field  approximates  that  of  the  human  mouth  to  the  extent  that  the  loss  in  power 
delivered  by  a  microphone  with  increasing  distance  between  the  mouth  and  the  micro- 
phone is  about  the  same  for  the  artificial  as  for  the  human  voice. 

TEE  VOCODER.  A  means  is  described  by  Dudley  for  continuously  analyzing  speech 
and  utilizing  the  results  of  the  analysis  to  synthesize  or  remake  the  speech.  It  is  based  on 
the  principle  that  the  intelligibility  of  speech  is  carried  by  the  relatively  low-frequency 
components  produced  by  cavity  modulation  and  that  the  "buzz  tone"  (vocal-cord  modula- 
tion) and  the  ''hiss  tone"  (frictions!  modulation)  simply  act  as  carrier  waves  which  are 
modulated  by  the  vocal  cavities.  Frequency  range  reduction  is  achieved  by  transmitting 
only  the  low-frequency  modulations  and  employing  them  to  modulate  locally  generated 
buzz  and  hiss  tones.  In  one  variation  called  the  voder,  speech  is  produced  artificially 
by  using  a  keyboard  manipulated  with  the  fingers  for  generating  the  carrier  waves  and  the 
low-frequency  modulations. 

VISIBLE  SPEECH.  A  development  described  by  Potter,  in  which  speech  is  continu- 
ously analyzed  and  the  results  of  the  analysis  portrayed  to  the  eye  in  the  form  of  visible 
patterns  that  one  can  learn  to  read,  holds  possibilities  in  the  fields  of  visual  hearing  and 
visual  telephony  for  the  deaf,  phonetic  printing  and  retranslation  into  sound,  the  selective 
operation  of  automatic  devices  by  voice  sounds,  and  in  the  specialized  fields  of  phonetics, 
linguistics,  foreign  language,  music,  etc  To  obtain  readable  patterns  of  speech  sounds, 
the  portrayal  emphasizes  the  modulations  that  are  important  to  intelligibility  as  illus- 
trated in  Fig.  1.  Frequency  is  shown  vertically,  time  horizontally,  and  intensity  by  shades 
of  gray.  The  stop  gap  (start-stop  modulation) ,  the  plosive  release  (frictional  modulation) , 
and  the  voice  bar  (vocal-cord)  modulation  are  indicated  for  the  voiced  stop  consonant  b. 
Likewise  the  vocal  resonance  bars  (cavity  modulation)  are  indicated  for  the  vowel  e.  The 
u  shows  only  two  resonance  bars  which  differ  in  position  from  those  of  the  e.  The  transi- 
tion from  e  to  u  is  shown  also.  The  fricative  fill  for  /  is  an  example  of  frictional  modula- 
tion. The  patterns  for  voiced  sounds  show  regular  vertical  striations,  the  space  between 


12-22 


ACOUSTICS 


the  striations  indicating  the  pitch.  Note  the  drop  in  pitch  toward  the  end  of  ^fiye." 
The  unvoiced  sounds  (frictional  modulation)  show  irregularly  spaced  vertical  striations 
indicating  a  lack  of  pitch.  The  patterns  are  read  from  the  characteristic  modulations  of 
the  sounds  and  the  transitions  between  them. 


Voice  bar  for  "8" 
_Stop  gap  for  "W 

Plosive  release  for  "B" 


IflosTve  release  for  "T" 

Fricative  fill  for 


~r 

0.6  0.8 

Time  in  seconds 
B       E  U       P  A  T  F  1 

FIG.  1.     Visible  Patterns  of  the  Words  "Be  up  at  Five" 


10.  SPEECH  POWER 

Since  the  wave  forms  of  speech  are  complex,  different  types  of  speech  power  have  been 
defined  as  follows: 

Instantaneous.  The  rate  at  which  sound  energy  is  being  radiated  by  the  speaker  at 
any  given  instant. 

Average.  The  average  speech  power  for  any  given  time  interval  is  the  average  value 
of  the  instantaneous  speech  power  over  that  interval. 

Peak.  The  maximum  value  of  the  instantaneous  speech  powers  occurring  in  a  given 
time  interval. 

Phonetic.  The  phonetic  speech  power  is  the  maximum  value  of  the  average  speech 
power  in  0.01-sec  intervals  of  a  vowel  or  consonant  sound.  It  is  the  maximum  value  of 
the  envelope  that  results  from  plotting  average  power  for  0.01-sec  intervals  against  time, 
as  the  sound  grows,  remains  steady,  and  decays. 

The  quantity  usually  measured  is  the  speech  pressure  at  some  convenient  distance  and 
direction  from  the  mouth.  Dunn  and  White  have  reported  the  results  of  measurements 
of  various  types  of  speech  pressure  and  comparisons  with  earlier  measurements,  and  Dunn 
and  Farnsworth  have  reported  on  the  directional  characteristics  of  the  mouth  as  a  radiator. 
The  results  are  summarized  in  Table  2.  They  comprise  statistical  measurements  of  speech 
pressure  vs.  frequency  range  for  conversational  speech  made  with  an  arrangement  for 
introducing  any  one  of  14  band-pass  niters  into  a  speech  circuit.  The  frequency  range 
below  500  cycles  was  covered  in  one-octave  steps.  Above  500  cycles,  each  filter  passed  a 
range  of  about  1/2  octave.  Provisions  were  made  for  measuring  instantaneous,  peak,  and 
rms  pressures  in  alternate  time  intervals  of  Vs  sec  and  also,  for  certain  cases,  in  time 
intervals  of  15  sec.  Some  600  intervals  of  Vs  sec  were  usually  measured  for  a  given  con- 
dition representing  an  integrated  time  of  75  sec. 

From  the  results  it  is  possible  to  calculate  the  various  types  of  speech  power.  This  is 
done  by  converting  pressure  to  power  per  square  centimeter  (intensity)  by  the  relation 
jf  =  p2/415,  where  I  =  intensity  in  microwatts  per  sq  cm  and  p  =  pressure  in  dynes  per 
sq  cm  at  a  given  distance  and  direction  from  the  mouth.  The  intensities  in  other  direc- 
tions are  obtained  by  weighting  in  accordance  with  the  directional  characteristics  of  the 
mouth,  and  P,  the  total  power  radiated,  is  then  obtained  by  integrating  over  a  spherical 
surface  having  a  radius  equal  to  the  distance.  The  results  of  such  calculations  are  given 
in  the  first  row  of  Table  2  in  the  form  of  ratios  expressed  in  decibels.  The  ratios  depend 
upon  the  distance  and  direction  from  the  mouth  and  the  frequency  range  in  which  p  is 
measured.  The  reference  position  is  designated  as  30,  0°,  0°,  signifying  30  cm  from  the 
mouth  at  zero  azimuth  and  altitude  angles,  i.e.,  directly  in  front  of  the  mouth.  The  values 
when  added  to  p  expressed  in  decibels  from  1  dyne  per  sq  cm  give  P  expressed  in  decibels 
from  1  microwatt.  The  values  are  based  on  distribution  measurements  for  one  voice. 


SPEECH  POWER 


12-23 


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12-24  ACOUSTICS 

The  second  and  third  rows  give  the  long  interval  rms  pressure  pat  30,  0°,  0°  in  decibels 
from  1  dyne  per  sq  cm.  The  rows  designated  "men"  and  "women  are  averages  lor  b  male 
and  5  female  speakers,  respectively.  By  long  interval  is  meant  a  time  average  over  some 
600  i/s-sec  intervals  or  75  sec.  Corresponding  values  in  other  bands  may  be  obtained  by 
taking  p2  as  proportional  to  band  width  in  cycles  per  second.  Rows  4,  5,  6,  and  7  show 
pressures  p  at  different  directions  relative  to  the  reference  position.  The  mouth  radiates 
maximally  in  a  direction  about  45°  down  from  the  horizontal.  Rows  8  and  9  show  peak 
pressures  that  are  exceeded  in  1  and  10  per  cent  of  the  i/g-sec  intervals  relative  to  the 
long-interval  values.  One  per  cent  peaks  more  than  20  db  above  long-interval  pressures 
occur  frequently.  Corresponding  values  for  other  bands  may  be  obtained  approximately 
for  peaks  occurring  less  than  10  per  cent  of  the  time,  by  proceeding  as  though  they  were 
rms  pressures.  The  peak-factor  data  are  essentially  the  same  for  both  men's  and  women's 
speech. 

Rows  10,  11,  and  12  show  the  distribution  of  i/s-sec  rms  pressures  relative  to  the  long- 
interval  values  given  in  rows  2  and  3.  Distributions  for  the  three  low-frequency  bands 
are  very  similar  to  the  whole  spectrum  distribution,  row  11.  Distributions  for  the  remain- 
ing bands  are  very  similar  to  that  for  the  700  to  1000  cycle  band  given  in  row  12.  These 
distributions  are  about  the  same  for  both  men's  and  women's  speech. 

Rows  13,  14,  and  15  give  distributions  of  instantaneous  and  peak  pressures  relative  to 
the  long-interval  rms  pressure.  The  distribution  for  instantaneous  pressures  is  based  on 
data  for  a  single  male  voice. 

The  last  two  rows  give  data  on  the  distribution  of  long-interval  rms  pressures  among 
various  speakers  reported  by  Fletcher. 

The  results  in  Table  2  indicate  a  long-interval  total  speech  power,  averaged  for  6  men, 
of  34  microwatts.  The  corresponding  figure  averaged  for  5  women  is  18  microwatts. 
These  values  are  somewhat  larger  than  that  of  10  microwatts  given  formerly.  Some  2  db 
of  this  difference  is  due  to  the  method  of  converting  measured  pressures  to  radiated  powers. 
These  values  obtain  for  continuous  connected  speech.  If  the  silent  intervals  (about  1/5 
to  1/3  of  the  total  time)  are  excluded,  the  average  is  increased  about  25  to  50  per  cent. 
The  above  values  hold  for  about  30  per  cent  of  the  speakers.  One  per  cent  of  the  speakers 
may  radiate  powers  in  excess  of  272  microwatts.  If  one  shouts  as  loudly  as  possible, 
the  total  power  may  reach  3400  microwatts,  an  increase  of  20  db.  For  a  very  faint  but 
intelligible  whisper,  the  total  power  may  fall  to  0.0034  microwatt,  a  drop  of  40  db. 

In  1  per  cent  of  i/s-sec  intervals,  the  peak  power  may  exceed  the  long-interval  total 
power  in  connected  speech  by  20  db.  Thus  total  peak  powers  of  the  order  of  3400  micro- 
watts may  be  reached  by  average  male  voices. 

The  ah  (father)  is  about  the  loudest  sound  in  speech.  Fletcher  reports  a  total  phonetic 
power  of  41  microwatts  for  this  sound.  Corresponding  values  for  other  sounds  may  be 
obtained  from  the  column  designated  Ll  of  Table  1.  These  obtain  for  discrete  words  or 
syllables  spoken  at  conversational  level.  Peak  powers  of  the  order  of  1600  microwatts 
for  the  ah  sound  were  obtained  under  these  conditions.  Wolf,  Stanley,  and  Sette  report 
total  phonetic  powers  of  1  watt  when  the  vowel  ah  is  sung  by  professional  singers.  To  be 
consistent  with  the  data  reported  in  Table  2,  the  total  phonetic  powers  given  above  should 
be  increased  by  2  db. 

Sivian  reports  on  changes  in  the  frequency  distribution  of  speech  power  as  a  speaker 
changes  from  a  low  (confidential)  talking  level  to  a  normal  (conversational)  level  and  to  a 
high  (declamatory)  level.  The  total  change  in  power  from  low  to  high  was  about  24  db. 
In  general,  power  .was  transferred  from  the  frequency  range  below  500  cycles  to  the  range 
between  500  and  4000  cycles  as  the  talking  level  was  raised.  To  be  representative  of 
declamatory  speech,  the  values  in  the  bands  below  500  cycles  given  in  row  2,  Table  2, 
should  be  decreased  about  4  db  and  values  between  500  and  4000  cycles  should  be  in- 
creased about  3  db. 

11.  POWERS  PRODUCED  BY  MUSICAL  INSTRUMENTS 

In  music,  as  in  speech,  various  aspects  of  power — peak,  average,  etc. — must  be  dealt 
with.  A  comprehensive  report  on  the  powers  produced  by  the  various  musical  instru- 
ments and  by  orchestras  has  been  given  by  Sivian,  Dunn,  and  White.  The  report  also 
describes  in  some  detail  the  band-pass  filter  apparatus  discussed  in  article  10  on  speech 
power.  Two  types  of  measurements  were  made  on  the  waves,  the  average  pressure 
in  15-sec  intervals  and  the  peak  pressures  in  i/s-sec  intervals.  The  pressures  are  the  field 
pressures  at  the  position  in  the  field  where  the  microphone  of  the  measuring  circuit  was 
placed.  For  the  measurements  on  individual  instruments,  the  position  of  the  microphone 
with  respect  to  the  source  varied  with  instrument.  For  the  orchestra,  the  microphone 


POWERS  PRODUCED  BY  MUSICAL  INSTRUMENTS      12-25 


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cylinder  having  drum  diameter. 

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8.5  db  for  l-ft  distance.  Radiati 
hemisphere. 

3-ft  distance.  Peak  pressure  incre 
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12-26 


ACOUSTICS 


was  placed  near  the  conductor's  stand.  By  making  assumptions  as  to  the  radiating  prop- 
erties of  the  instruments,  and  of  the  orchestra,  their  total  power  outputs  could  be  estimated. 
Table  3  gives  the  average  and  peak  pressures  at  the  positions  of  measurement  as  indi- 
cated. The  columns  "total  peak  power"  give  the  total  power  radiated  by  the  instru- 
ments computed  on  the  basis  of  assumptions  indicated  in  the  table.  The  left  half  of  the 
table  applies  to  the  whole  spectrum;  the  right  half  to  the  bands  containing  the  maximum 

***  An  'average  of  three  methods  was  used  for  determining  the  total  power  radiated  by  the 
piano.  In  the  first  method,  the  diffuse  pressure  in  a  room  of  known  reverberation  was 
measured  and  the  power  calculated.  In  the  second  method,  from  auxiliary  measurements, 
it  was  shown  that  the  measured  diffuse  pressure  was  the  same  as  if  the  total  power  output 
were  distributed  over  a  sphere  of  radius  5.37  ft.  In  the  third  method,  the  diffuse  pressure 
was  compared  with  the  pressure  measured  in  the  opening  between  the  raised  top  (grand 
piano)  and  the  sounding  board.  The  radiated  power  was  assumed  to  be  uniform  over 
the  area  of  the  opening,  3  ft  by  6  ft.  The  three  methods  gave  peak  powers  of  0.166,  0.437, 
and  0.198  watt,  respectively. 

The  measurements  on  the  15-piece  orchestra  were  made  in  the  same  room  as  the  piano 
measurements,  and  methods  1  and  2  were  used  to  obtain  the  total  power.  The  sounds 
from  the  75-piece  orchestra  were  picked  up  in  a  theater,  and  suitable  acoustic  data  for 
using  the  above  methods  were  not  available.  It  was  assumed  that  the  radiation  was 
uniform  over  a  hemisphere  of  15-ft  radius.  The  measurements  on  the  pipe  organ  were 
made  in  the  same  theater. 

Because  of  the  uncertain  assumptions  in  calculating  total  powers,  the  measured  values 
of  pressure  are  also  given  in  Table  3.  The  peak  pressures  were  measured  within  a  range 
of  db3  db,  and  the  percentage  of  i/s-sec  intervals  that  the  power  fell  within  this  range  is 
indicated.  The  peak  pressures  are  useful  in  determining  the  amplitudes  that  pick-up 
instruments  and  amplifiers  must  handle,  and  the  total  peak  powers  for  determining  the 
requirements  that  must  be  met  by  power  amplifiers  and  loudspeakers  when  reproduction 
of  the  original  level  is  desired.  The  most  powerful  single  instrument  is  a  bass  drum,  which 
radiates  a  power  of  about  25  watts,  primarily  in  the  low-frequency  ranges.  A  large  orches- 
tra is  capable  of  radiating  power  of  about  60  to  70  watts.  This  is  about  70,000  times  the 
peak  power  of  speech. 

Figure  2  (from  Fletcher)  shows  the  frequency  distribution  of  the  maximum  and  most 
probable  peak  powers  for  a  75-piece  orchestra.  The  curves  are  based  on  average  measure- 
ments of  four  selections  which  gave  whole-spectrum  peak  powers  from  8  to  66  watts,  and 


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2.7      65.4        130.8       261.2       523.2       1046        2092        4185        8370 
Frequency  in  Cycles  per  Second 
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Frequency  Level  in  Octaves  from  16.35  Cycles 

FIG.  2.     Maxim  mm  and  Most  Probable  Peak  Powers 

average  powers  from  0.08  to  0.13  watt.  The  zero  line  corresponds  to  an  average  power 
of  about  1/io  "watt.  The  ordinate  is  the  ratio  expressed  in  decibels  of  the  peak  power  per 
octave  to  the  whole-spectrum  average  power. 

A  violin  player,  when  asked  to  play  at  the  lowest  level  that  could  be  used  with  an 


TESTS  OF  SPEECH  AND  MUSIC  TRANSMISSION       12-27 

audience,  produced  a  whole-spectrum  average  pressure  of  0.52  dyne  per  sq  cm  for  a  3-ft 
distance.  The  highest  peak  observed  for  the  bass  drum  was  1250  dynes  per  sq  cm.  Thus 
a  range  _of  at  least  68  db  is  indicated  between  the  highest  and  lowest  amplitudes  in  music. 
The  estimated  range  probably  errs  in  the  direction  of  being  too  small  rather  than  too  large. 

12.  TESTS  OF  SPEECH  AND  MUSIC  TRANSMISSION 

The  performance  of  a  system  for  the  transmission  and  reproduction  of  speech  or  music 
can  be  expressed  in  terms  of  objective  measurements  involving  a  determination  of  its 
characteristics  for  steady-state  sinusoidal  waves  of  different  frequency,  or  in  terms  of 
subjective  measurements  involving  the  satisfaetoriness  of  the  system  from  the  viewpoint 
of  the  person  receiving  the  signals.  This  section  is  concerned  with  the  subjective  type  of 
measurement  which  involves,  for  speech,  the  recognizability  and  the  naturalness  of  the 
reproduced  sounds,  and  for  music,  the  emotional  and  esthetic  properties  of  the  reproduc- 
tion. Measurements  of  this  kind  may  be  classed  as  laboratory  and  field  measurements. 
The  former  are  particularly  adapted  to  fundamental  studies  of  the  effects  of  various 
physical  factors  on  the  reproduction;  the  latter,  to  evaluating  performance  under  condi- 
tions of  actual  use. 

LABORATORY  TESTS.  The  articulation  test  has  come  into  wide  use  as  a  laboratory 
method  of  measuring  the  recognizability  of  received  speech  sounds.  One  form  of  the  test 
which  was  used  in  obtaining  the  data  given  in  the  next  article  has  been  described  by 
Fletcher  and  Steinberg.  In  this  form,  a  speaker  utters  the  various  speech  sounds  and  an 
observer  writes  down  the  sounds  which  he  hears.  The  observed  sounds  are  compared 
with  the  called  sounds,  and  the  percentage  of  the  called  sounds  that  were  correctly  recog- 
nized is  obtained.  This  percentage  is  the  articulation.  To  call  the  sounds,  they  are 
combined  in  a  random  manner  into  syllables  of  the  oonsonant-vowel-consonant  type 
which  have  no  meaning  in  English.  The  syllables  are  spoken  as  parts  of  introductory 

sentences,  such  as,  "Please  record  the  syllable ."  The  sounds  used  are  those  shown  in 

Table  1  (p.  12-20) ,  and  they  occur  with  uniform  frequency  in  the  testing  lists.  The  per- 
centage of  the  total  number  of  spoken  syllables  which  are  correctly  observed  is  called 
the  "syllable  articulation."  The  "sound  articulation"  is  the  percentage  of  the  total  num- 
ber of  spoken  sounds  which  are  correctly  observed.  When  attention  is  directed  toward 
a  specific  fundamental  sound,  e.g.,  b,  the  term  "individual  sound  articulation"  is  used. 
Similarly,  "vowel  articulation"  is  the  percentage  of  the  total  number  of  spoken  vowel 
sounds  which  are  correctly  observed. 

To  obtain  reproducible  and  representative  results  it  is  important  that  the  testing  crew 
have  normal  voices  and  hearing  and  that  they  be  thoroughly  familiar  -vrith  the  method. 
The  best  number  of  voices  and  observers  and  the  amount  of  testing  material  required 
depend  on  the  discrimination  which  is  desired.  For  most  of  the  data  reported  in  the  next 
article,  each  of  8  callers  reads  a  list  of  66  syllables  to  4  observers.  Practice  effects  of  two 
kinds  occur,  a  short-term  period  of  improvement  when  the  crew  tests  unfamiliar  condi- 
tions, and  gradual  long-term  changes  in  skill.  The  first  may  be  eliminated  by  repeating 
tests;  the  second  may  be  evaluated  by  suitable  control  tests  on  reference  circuits.  The 
use  of  automatic  equipment  to  facilitate  the  control  of  variable  factors  is  very  advanta- 
geous, particularly  when  considerable  testing  is  done.  Castner  and  Carter  have  described 
equipment  of  this  kind.  The  sounds  that  the  observer  hears  are  recorded,  and  the  re- 
corded result  is  analyzed  into  various  articulation  values,  automatically  within  a  few 
seconds  after  the  test  is  completed,  by  suitable  selecting  and  recording  equipment.  In 
addition,  various  circuit  conditions  to  be  tested  are  set  up  at  the  proper  time,  the  talking 
levels  of  the  callers  are  measured,  and  numerous  other  steps  in  a  test  are  accomplished 
automatically. 

Another  type  of  test  that  has  been  used  somewhat  is  the  "intelligibility  test."  In  some 
such  tests,  short  sentences  of  the  interrogative  or  imperative  form  containing  a  simple 
idea  are  used.  The  sentences  are  considered  to  be  understood  if  the  observer  either 
records  the  sentence  or  records  an  intelligent  answer.  In  others,  English  words,  selected 
at  random  from  print,  are  used.  The  "discrete  sentence  intelligibility"  is  the  percentage 
of  the  total  number  of  called  sentences  which  are  correctly  understood.  Similarly,  the 
"discrete  word  intelligibility"  is  the  percentage  of  called  words  correctly  observed.  Fig- 
ures 3  and  4  illustrate  the  types  of  relations  between  these  intelligibilities  and  syllable 
articulation.  The  intelligibility  tests  are  useful  in  testing  very  poor  systems  having  an 
articulation  of  only  a  few  per  cent,  which  are  used  sometimes  in  fundamental  studies. 

Articulation  and  intelligibility  tests  afford  a  quantitative  measure  of  the  intelligibility 
of  reproduced  speech.  The  naturalness  of  speech  and  music,  and  the  emotional  "and 
esthetic  appeal  of  musical  sounds,  are  less  definite,  and  have  been  studied  principally  by 


12-28 


ACOUSTICS 


means  of  the  AB  judgment  test.  In  this  test,  the  observer  listens  to  two  conditions  A 
and  B,  and  judges  which  condition  is  distorted,  or  which  condition  is  the  poorer  etc., 
depending  on  the  study.  Sometimes  one  condition  may  be  the  original  sound  and  the 


80 


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20 


20 


so 


100 


FIG.  3. 


40  60 

Syllable  Articulation 
Discrete  Sentence  Intelligibility  vs.  Syllable  Articulation 


other  condition  the  repro- 
duced sound.  Although  such 
tests  do  not  always  give  a 
quantitative  measure  of  the 
effects  of  distortion  on  natu- 
ralness, esthetic  appeal,  etc., 
they  do  afford  a  means  of 
measuring  the  just-noticeable 
amounts  of  distortion.  This 
type  of  test  was  used  in  stu- 
dies on  the  audible  frequency 
ranges  of  music,  speech,  and 
noise  which  have  been  re- 
ported by  Snow. 

FIELD  TESTS.  The  re- 
sults of  such  laboratory  tests 
cannot  be  taken  to  indicate 
directly  the  performance  of 
actual  circuits  under  service 
conditions.  In  service,  there 
is  a  wide  range  of  variation  in 
a  number  of  conditions,  such 
as  the  subject  matter  of  con- 
versations, the  manner  in 
which  the  transmitter  is  spo- 
ken into,  the  way  in  which  the 
receiver  is  held  to  the  ear,  and 
the  reactions  of  telephone  users  to  the  various  circuit  conditions.  Because  it  is  difficult 
to  specify  definitely  such  conditions  and  hence  simulate  them  in  the  laboratory,  the  repeti- 
tion test,  which  has  been  described  by  Martin,  has  come  into  use.  In  this  test,  observations 
are  made  on  telephone  circuits  under  service  conditions.  Essentially,  the  test  is  that  of 
noting  the  repetitions  in  a 
telephone  conversation  and  10° 
from  a  number  of  observa- 
tions determining  the  repe- 
tition rate.  Martin's  paper 
describes  repetition  obser- 
vations on  conversations  be- 
tween several  hundred  sta- 
tions in  the  American  Tele-  £ 
phone  and  Telegraph  Com-  ~ 
pany  headquarters  building  = 
and  a  similar  number  of  sta-  j= 
tions  in  the  Bell  Telephone  | 
Laboratories  building.  To  5 
provide  data  for  rating  *• 
transmission  performance,  J 
the  station  instruments  and  & 
the  transmission  character- 
istics of  the  interconnecting 
trunks  were  varied  and  the 
effects  on  repetition  rate  de- 
termined. McKown  and 
Emling  have  described  a 
system  of  effective  trans- 
mission data  for  rating  tele- 
phone circuits,  based  on  the 
results  of  such  tests  and 
articulation  tests 

OTHER  TESTS.     During  the  war  a  numbe 


P 


20 


40  60 

Syllable  Articulation 


FIG.  4.     Discrete  Word  Intelligibility  vs.  Syllable  Articulation 


of  new  tests  were  devised  for  testing 
speech  transmission  under  conditions  obtaining  in  military  operations.  See:  Transmission 
and  Reception  of  Sounds  under  Combat  Conditions,  Miller  and  Weiner,  Columbia  University 
Press,  New  York,  1948. 


EFFECTS    OF   DISTOETION   ON   SPEECH   AND   MUSIC       12-29 

PREDICTION  OF  ARTICULATION  TEST  RESULTS.  On  many  types  of  speech- 
transmission  circuits  it  is  possible  to  measure  the  circuit  characteristics  and  noise  levels 
and  by  means  of  these  data  compute  the  articulation  score  that  would  be  obtained.  Com- 
putational methods  are  also  useful  hi  the  design  of  equipment  for  speech  circuits  and  of 
considerable  theoretical  interest  in  the  study  of  the  interpretation  of  speech  sounds. 
See:  Beranek,  L.  L.,  Design  of  Speech  Communication  Systems,  Proc.  IRE,  Vol.  35,  SO 
(1947);  French  and  Steinberg,  Factors  Governing  Intelligibility  of  Speech  Sounds,  /. 
Acous.  Soc.  Am.,  Vol.  19,  90  (1947);  and  Fletcher  and  Gait,  The  Perception  of  Speech  and 
Its  Relation  to  Telephony,  /.  Acous.  Soc.  Am.,  July  1950. 

BIBLIOGRAPHY 

Bureau  of  Publications,  Bell  Telephone  Laboratories,  High  Speed  Motion  Pictures  of  the  Human 

Vocal  Cords.     (Two  reels  of  slow-motion  views  available  for  loan  on  request.) 
Castner  and  Carter,  Developments  in  the  Application  of  Articulation  Testing,  Bell  Sys.  Tech.  J.,  July 

1933,  p.  34 /. 

Crandall,  I.  B.,  Sounds  of  Speech,  Bell  Sys.  Tech.  J.,  October  1925,  p.  5S6. 
Dudley,  H.,  Remaking  Speech,  J.  Acous.  Soc.  Am.,  Vol.  11,  169  (October  1939). 
Dudley,  H.,  The  Carrier  Nature  of  Speech,  Bell  Sys.  Tech.  J.,  Vol.  19,  495  (October  1940). 
Dudley,  Homer,  and  Otto  O.  Gruenz,  Jr.,  Visible  Speech  Translators  with  External  Phosphors,  J. 

Acous.  Soc.  Am.,  Vol.  18,  62  (July  1946). 
Dunn,  H.  K.,  and  D.  W.  Farnsworth,  Exploration  of  Pressure  Field  Around  the  Human  Head  during 

Speech,  /.  Acous.  Soc.  Am.,  Vol.  10,  184  (January  1939). 
Dunn,  H.  K.,  and  S.  D.  White,  Statistical  Measurements  on  Conversational  Speech,  J,  Acous.  Soc. 

Am.,  Vol.  11,  278  (January  1940). 
Firestone,  F.  A.,  Artificial  Larynx  for  Speaking  and  Choral  Singing,  J.  Acous.  Soc.  Am.,  January  1940, 

p.  11. 

Fletcher,  H.,  Physical  Characteristics  of  Speech  and  Music,  Bdl  Sys.  Tech.  J.,  July  1931,  p.  349. 
Fletcher  and  Steinberg,  Articulation  Testing  Methods,  Sett  Sys.  Tech.  J.,  October  1929,  p.  806. 
French,  Carter,  and  Koenig,  Words  and  Sounds  of  Telephone  Conversation,  Bell  Sys.  Tech.  J.,  April 

1930,  p.  290. 
Koenig,  W.,  H.  K.  Dunn,  and  L.  Y.  Lacy,  Sound  Spectrograph,  J.  Acous.  Soc.  Am.,  Vol.  IS,  19  (July 

1946). 
Kopp,  G.  A.,  and  H.  C.  Green,  Basic  Phonetic  Principles  of  Visible  Speech.  J.  Acous.  Soc.  Am.,  Vol. 

18,  62  (July  1946). 

Lewis,  Don,  Vocal  Resonance,  J.  Acous.  Soc.  Am.,  VoL  8,  91  (October  1936). 
Lewis  and  TutMll,  Resonant  Frequencies  and  Damping  Constants,  J.  Acous.  Soc.  Am.,  VoL  11,  451 

(April  1940). 

Martin,  W.  H.,  Rating  the  Transmission  Performance  of  Telephone  Circuits,  BeU  Sys.  Tech.  J.,  Janu- 
ary 1931,  p.  116. 
McKown  and  Emling,  A  System  of  Effective  Transmission  Data  for  Rating  Telephone  Circuits,  BeU 

Sys.  Tech.  J.,  July  1933,  p.  331. 

Paget,  Sir  Richard,  Production  of  Artificial  Vowel  Sounds,  Proc.  Roy.  Soc.,  VoL  102,  752  (1923). 
Potter,  R.  K.,  Visible  Patterns  of  Sound,  Science,  VoL  102,  463  (Nov.  9,  1945). 
Potter,  R.  K.,  G.  A.  Kopp,  and  H.  C.  Green,  Visible  Speech.     Van  Nostrand  (1947). 
Riesz,  R.  R.,  An  Artificial  Larynx,  J.  Acous.  Soc.  Am.,  January  1930,  p.  273. 
Riesz,  R.  R.,  and  L.  Schott,  Visible  Speech  Cathode  Ray  Translator,  J.  Acous.  Soc.  Am.,  Vol.  18,  50 

(July  1946). 

Sacia  and  Beck,  Power  of  Fundamental  Speech  Sounds,  BeU  Sys.  Tech.  J.,  July  1926,  p.  393. 
Sivian,  L.  J.,  Speech  Power  and  Its  Measurement,  Btll  Sys.  Tech.  J.,  October  1929,  p.  646. 
Sivian,  Sunn,  and  White,  Absolute  Amplitudes  and  Spectra  of  Musical  Instruments  and  Orchestras, 

J.  Acous.  Soc.  Am.,  January  1931,  p.  330. 
Steinberg,  Application  of  Sound-measuring  Instruments  to  the  Study  of  Phonetic  Problems,  J.  Acous. 

Soc.  Am.,  July  1934,  p.  16.  _  , 

Steinberg,  J.  C.,  and  N.  R.  French,  The  Portrayal  of  Visible  Speech,  J.  Acous.  Soc.  Am.,  VoL  IS,  4 

(July  1946). 

Stewart,  J.  Q.,  An  Electrical  Analogue  of  the  Vocal  Organs,  Mature,  September  1922,  p.  311. 
Wolf,  Stanley,  and  Sette,  Quantitative  Study  of  the  Singing  Voice,  J.  Acous.  Soc.  Am.,  VoL  6,  255 

(April  1935). 


EFFECTS  OF  DISTORTION  ON  SPEECH  AND  MUSIC 

By  John  C.  Steinberg  and  W.  A.  Munson 

In  studies  of  the  effects  of  distortion,  it  is  desirable  to  have  a  suitable  reference  system. 
Under  certain  conditions,  air  transmission  from  speaker  to  observer  approaches  such  a 
reference.  Owing  to  practical  difficulties  in  changing  and  controlling  an  ah-  transmission 
system,  it  is  convenient  to  use  electrical  systems  composed  of  transmitters,  amplifiers, 
and  receivers  or  loudspeakers.  By  suitable  calibration  and  equalization,  such  systems 
can  be  made  to  approach  air  in  transmission  properties,  and  various  amounts  of  degrada- 
tion or  improvement  can  be  introduced.  Circuits  of  this  type  have  been  studied  by  means 
of  articulation  and  judgment  tests  and  the  effects  of  various  kinds  of  distortion  de- 
termined. 


12-30 


ACOUSTICS 


13.  EFFECT  OF  FREQUENCY  DISTORTION 

AUDIBLE  FREQUENCY  RANGES  OF  MUSIC,  SPEECH,  AND  NOISE.  One  step 
in  the  evaluation  of  the  effects  of  distortion  on  the  interpretation  of  sounds  is  the  deter- 
mination of  the  just-noticeable  amounts  of  distortion.  Snow  has  reported  the  results  of 
tests  on  the  lower  and  upper  frequency  limits  of  various  types  of  sounds.  The  sounds 
as  produced  in  one  room  were  picked  up  by  a  microphone  and  transmitted  to  a  second 
room  where  they  were  reproduced  by  means  of  loudspeakers.  By  means  of  filters,  all 
frequencies  above  or  below  any  desired  cutoff  could  be  suppressed.  By  comparing  the 
unfiltered  with  the  filtered  transmission,  trained  observers  determined  the  just-detectable 
upper  and  lower  cutoffs.  The  results  of  the  tests  are  shown  in  Fig.  1.  The  ends  of  the 

Audible  Frequency  Range 
for  Music  Speech  and  Noise 
-  Actual  Tone  Range 


nilWIIIIIltl  Accompanying  Noise  Range 
°- Cut-off  Frequency  of  Filter 
Detectable  in  80%  of  Tests 


Bass  Drurn- 

\ 

BflHW 

HUH 

p 

Jii 

Bass  Viol  

nit 

Ce|l0  . 

•(  HffHJWJWI 

Violin  

HIK 

nun 

Trombone 

F  ench  Ho 

n     o  n— 

in  n 

Bass  Saxophone  

it 

laiiiiiiiui 

Bass  CtaE&Lfit  

Clarinet  

j        J 

Sopramo  Swfophofle— 
Oboe  

'      " 

' 

Flute  

Male  Speech-  - 

Female  Speech  

°" 

Key  JiQgUng-—  —     ——  — 

0 

4 

0 

100                        500      1,000                     5,000  10,000  20, 

Frequency  In  Cycles  per  Second 

4  5  6  7  8  9  10 

Frequency  Level  in  Octaves  from  16.35  Cycles 

FIG.  1.    Audible  Frequency  Ranges 

lines  representing  the  audible  ranges  are  the  cutoffs  where  the  correct  judgments  amounted 
to  60  per  cent.  Circles  mark  the  cutoffs  where  80  per  cent  correct  judgments  were  ob- 
tained The  ^  musical  sounds  produced  by  many  of  the  musical  instruments  were  accom- 
panied by  noise  such  as  key  clicks,  lip  noises,  "buzz"  of  reeds,  and  hissing  of  air.  As  far  as 
possible,  the  observers  tried  to  distinguish  between  tone  and  noise.  The  lines  indicate 
their  decision  as  to  the  actual  tone  range,  and  the  short  vertical  lines  indicate  the  noise 
range.  In  all  cases,  the  upper  frequency  limit  exceeds  very  greatly  the  frequencies  of  the 
Highest  notes  that  the  instruments  were  capable  of  producing.  Except  for  the  piano  the 

^ 


ad 

Uonal  type  of  test  was  made  with  music  furnished  by  an  18-pieoe  orchestra.  In  this  test 
the  filtered  reproduction  was  always  presented  on  the  B  condition,  and  the  unfiltered  on 
the  A  condition.  The  observers  were  asked  to  rate  the  quality  of  the  B  condition  as  a 
percentage  of  the  A  condition.  A  ratmg  of  less  than  100  per  cent  indicated  a  degradation; 
Lrr^l  th*T  th*JV100.Per  ce°*  *******  an  improvement.  The  various  Sitoffs  oc^ 
cm-red  on  the  ,  B  condition  m  random  order.  The  results  of  this  test  are  given  in  Fig  2 
The  judgment  of  the  ten  observers  taking  part  in  this  test  was  that  the  quality  ^creased 
rapldly  as  the  cutoff  was  extended  upward  to  8000  cycles  or  downward  to  80  cycles 


AKTICULATION  TESTS 


12-31 


The  results,  in  view  of  the  nature  of  the  test,  were  surprisingly  consistent.  In  only  about 
3  per  cent  of  some  400  observations  was  the  quality  of  a  filtered  system  given  a  rating 
greater  than  100  per  cent,  so  that,  in  general,  the  reproduction  of  the  full  audible  range 
was  preferred. 


20 


90 
80 

n 

3 

High  Pass  Filters 
n  First  Test  (Blue  Danube) 
o  Second  Test  (In  a  Village) 

1      \    \ 

&T_ 

i 

ill 

i^° 

70 
^60 
J-50 

CO 

0-40 
30 
20 
10 

x 

Low  Pas: 
&  Firs 
c  Sec 

niters 
tTest 

and  Test 

I/ 

O.r 

\ 

/J 

\ 

i 

i 

^ 

i 

\ 

. 

i 

\ 

/ 

'j 

A 

\ 

s    1 

r^ 

j 

1ft* 

O. 

| 

50        100 


500 


Cut-off  Frequency  :n  Cycles  per  Second 
FIG.  2.     Orchestral  Quality  vs.  Cut-off  Frequency 


5000   100OO  200QO 


14.  ARTICULATION  TESTS 

RELATION  BETWEEN  ARTICULATION  AND  LEVEL  OF  SPEECH  ABOVE 
THRESHOLD.  The  articulation  test  affords  a  means  of  dealing  with  the  effects  of  dis- 
tortion in  a  more  quantitative  way  than  does  the  judgment  test.  An  extensive  series  of 
articulation  studies  have  been  reported  by  Fletcher  and  Steinberg,  for  widely  varying 
types  of  distortion.  Figure  3  shows  the  effect,  on  syllable  articulation,  of  changing  the 
received  speech  level  (see  also  article  31).  In  the  range  from  10  to  40  db  above  threshold, 
the  articulation  increases  rapidly  with  increasing  level.  In  the  range  from  50  to  90  db 
there  is  little  change  with  increasing  level.  Experimentally  it  has  been  determined  that, 
if  a  speaker  radiates  the  average  speech  power  of  10  microwatts,  a  listener  of  normal  hear- 


100 


"r 


20 


40  60 

DB  above  Threshold 


FIG.  3.    Effects  of  Speech  Level  on  Syllable  Articulation 

ing  1  meter  away  in  the  undisturbed  sound  field  will  "hear  the  speech  at  a  level  of  about 
62  db  above  threshold.  Figure  3'  shows  that  this  level  is  sufficient  to  insure  very  good 
articulation.  It  is  a  level  that  would  obtain  in  conversing  at  a  meter's  distance  in  a  cpiiet, 
well-damped  room.  A  more  detailed  picture  of  the  effects  of  speech  level  on  articulation 


12-32 


ACOUSTICS 


60 


40 


20 


.  Phonetic  Power 
DB  from  lO^Watt 
-  Long  Vowels 113 

—  Short  Vowels 113 

—  Nasal  Cons. 106 

Stop  Cons. 98 

Fricative  Cons. 99 


is  obtained  by  grouping  the  sounds  into  five  groups,  where  the  sounds  in  each  group  have 
somewhat  similar  properties.    The  grouping  used  is:  long  vowels — a,  a,  e,  o,  o',  u;  short 

vowels — a',  e,  i,  o,  u,  w,  y;  nasal 
consonants — I,  m,  n,  ng,  r\  stop  con- 
sonants— 6,  ch,  d,  g,  h,  j1,  k,  p,  t; 
fricative  consonants—/,  s,  sh,  $t,  th, 
thf  (then),  v,  z,  zh  (see  Table  1,  p. 
12-20).  Figure  4  shows  the  group 
articulation  vs.  speech  level.  As 
the  level  is  increased  from  thresh- 
old, the  vowels  are  the  first  to  be 
recognized  and  are  followed  in  order 
by  the  nasal,  stop,  and  fricative 
consonants.  The  apparent  differ- 
ences between  the  levels  of  the 
sound  groups,  as  judged  from  artic- 
ulation tests,  correspond  approxi- 
mately with  the  differences  in  av- 
erage phonetic  powers  shown  in 
the  insert.  For  example,  the  vowel 
articulation  at  a  level  of  25  db  is 
about  90  per  cent.  The  nasal  con- 
sonant group  must  be  increased  6 
db  from  the  25-db  level  for  an  ar- 
ticulation of  90  per  cent,  and  the 
stop  and  fricative  consonants  must 
FIG.  4.  Effects  of  Speech  Level  on  Sound  Articulation  be  increased  by  10  and  13  db,  re- 
spectively. 

EFFECT  ON  ARTICULATION  OF  REMOVING  PORTIONS  OF  THE  SPEECH 
FREQUENCY  RANGE.  To  determine  the  effects  of  frequency  distortion  on  articula- 
tion, electrical  filters  were  introduced  into  a  circuit  that  was  otherwise  reasonably  free 
from  distortion.  One  type  of  filter  (low-pass  filter)  eliminated  all  components  above  a 
chosen  cutoff  frequency  and  uniformly  transmitted  the  remainder  of  the  range.  Another 
type  (high-pass  filter)  eliminated  all  components  below  a  chosen  cutoff  frequency.  Fig- 
ure 5  shows  the  effects  on  articulation  of  changing  the  cutoff  frequency  of  these  filters. 
The  level  of  the  speech,  before  the  filter  was  introduced,  was  held  constant  at  70  db  above 
threshold.  The  curves  marked  L.P.  are  for  the  case  where  components  below  the  cutoff 
frequency  were  transmitted;  for  the  H.P.  curves,  components  above  the  cutoff  frequency 
were  transmitted.  The  cutoff  frequency  at  which  the  L.P.  andr  H.P.  curves  intersect  is 


/v 


20 


40  60 

DB  above  Threshold 


100 


H 

P. 

"""""^ 

^ 

LJP. 

/ 

C 

Stop 
ons.A 

rf. 

II 

HJ 

P- 

*7f 

L.P. 

It 

/ 

/ 

/ 

Fricat'i 
Cons.  A 

/e 

1 

1Q2  2     4  68io3  2     4  68  4    ~22~ 

Cut-off  Frequency 
FIG.  5.     Articulation  vs.  Cut-off  Frequency 

a  frequency  such  that  the  frequency  range  below  it  has  an  articulation  equal  to  the  fre- 
quency range  above  it.  This  intersection  frequency  is  always  higher  for  female  than  for 
male  speech,  indicating  that  the  higher-frequency  ranges  are  more  important  for  female 
speech.  The  frequency  range  in  which  the  curves  bend  down  is  the  range  of  importance 
in  articulation,  which  as  may  be  seen  varies  from  one  sound  group  to  another  P 


ABTICULATION)  TESTS 


12-33 


^  EFFECTS  OF  AN  EXTRANEOUS  NOISE  ON  ARTICULATION.  As  pointed  out 
in  the  article  on  masking,  noise  has  the  effect  of  shifting  the  threshold  of  audibility  of 
other  sounds  heard  in  the  presence  of  noise.  If  the  masking  of  a  noise  is  reasonably 
uniform  over  the  important  speech  frequency  range,  articulation  tests  made  in  the  presence 

°^  nCf^j    °^  the  main  eSect  *  to  shift  the  curves  of  articulation  vs.  level  above 

threshold.  Figure  6  shows  articulation  vs.  level  for  a  quiet  circuit  and  for  the  circuit  with 
noise  present  at  three  different  levels.  The  corresponding  masking  effects  of  the  noise 
are  shown  by  the  insert  (noise  audiogram).  The  abscissa  of  the  articulation  vs,  level 
curves  gives  the  decibels  above  threshold  of  the  speech  on  a  quiet  circuit.  The  amounts 
of  the  shift  in  the  articulation  curves  correspond  reasonably  well  with  the  average  amounts 
of  masking  produced  by  the  noise.  When  both  the  speech  and  noise  levels  are  high,  the 


100 


Eflect  of  Noise  on  Articulation 


20 


40 


100 


50  80 

DB  Above  Threshold 

FIG.  6.    Effects  of  Noise  on  Articulation* 


observed  articulation  values  are  less  than  would  be  indicated  by  a  simple  shift  in  the 
articulation-level  curves.  Modulation  between  the  speech  and  the  noise  then  takes  place 
in  the  ear,  which  decreases  the  maximum  value  of  articulation  that  can  be  obtained. 
When  the  masking  is  confined  to  a  limited  portion  of  the  speech  frequency  range,  such  as 
that  produced  by  a  2000-cycle  tone,  the  effects  on  articulation  cannot  be  represented  by  a 
simple  shift  in  the  articulation  vs.  level  curves.  In  this  ease,  the  curves  are  both  shifted 
and  changed  in  shape.  Moderate  amounts  of  deafness  affect  articulation  in  much  the 
same  way  as  noise.  When  the  hearing  loss  is  uniform  with  frequency,  the  articulation 
to  be  expected  is  that  obtained  by  a  corresponding  shift  in  the  normal  articulation  vs. 
level  curve. 

EFFECT  OF  RESONANCE  TYPE  OF  FREQUENCY  DISTORTION  ON  ARTICU- 
LATION. A  very  common  type  of  distortion  occurring  in  transmission  systems  is  that 
of  resonance.  Figure  7  shows  the  effects  of  a  resonant  peak  at  1100  cycles,  on  articulation. 
The  left-hand  eurves,  numbered  2  and  3,  show  the  loss  in  decibels  at  each  frequency 
relative  to  the  loss  in  the  uniform  system  designated  as  1.  The  right-hand  curves  show 
articulation  vs.  level  for  the  three  systems.  The  abscissa  is  the  decibels  above  threshold 
of  the  normal  speech,  i.e.,  before  inserting  the  resonance  distortion.  At  the  lower  speech 
levels  the  decrease  in  articulation  due  to  the  distortion  is  much  greater.  Most  of  the 
articulation  loss  for  a  resonance  type  of  distortion  appears  to  arise  because  those  sounds 
which  have  then-  important  components  in  the  frequency  ranges  removed  from  resonance 
axe  received  at  effectively  lower  than  normal  levels.  For  example,  a  large  part  of  the 
important  frequency  range  for  fricative  consonants  is  received  at  a  level  some  30  to  40  db 
below  normal  in  curve  3.  When  the  normal  level  is  low,  this  further  reduction  causes 
appreciable  articulation  losses.  Although  the  articulation  at  optimum  received  levels  may 
not  b£  seriously  affefrt&l  by  resonance  distortion*  the  tonal  character  or  naturalness  of  tfee 
distorted  sp^ecfe  is  appreciably  impaired. 


12-34 

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40 

50 

1 

/ 

\ 

/ 

)> 

\ 

/ 

,s 

/ 

' 

\ 

/ 

^" 

s^ 

s. 

k 

s 

S. 

2         345789,             2        34567  89     4             40                 50               S( 
O2                                                   103                                                  104         DB  above  Threshold 
Frequency                                                           of  Normal  Speech 

FIG.  7.     Effects  of  Resonance  Type  of  Frequency  Distortion  on  Articulation 

EFFECTS  OF  NON-LINEAR  DISTORTION  ON  ARTICULATION.  Non-linear  dis- 
tortion arises  when  the  output  of  a  system  is  not  proportional  to  the  input.  There  are 
two  cases  of  interest:  (a)  when  the  output  increases  at  a  more  rapid  rate  than  the  input; 
(6)  when  the  output  increases  at  a  slower  rate  than  the  input.  In  both,  modulation 
products  or  extraneous  frequencies  are  introduced  for  the  intense  sounds  when  they  cover 
an  appreciable  part  of  the  curved  portion  of  the  input  vs.  output  characteristic.  The 
result  is  a  decrease  in  the  articulation  of  such  sounds.  The  amount  of  the  decrease  de- 
pends upon  the  amount  of  the  curvature.  In  addition,  in  (a)  the  intense  sounds  are 
amplified  relative  to  the  faint  sounds,  which  has  the  effect  of  decreasing  the  articulation 
of  the  faint  sounds.  In  (b)  the  faint  sounds  are  amplified  relative  to  the  intense  sounds. 
This  has  the  effect  of  increasing  the  articulation  and  may,  if  the  received  speech  level  is 
below  the  optimum  receiving  level,  offset  the  decrease  due  to  modulation  effects.  For 
optimum  receiving  levels,  however,  the  net  result  is  always  a  decrease  in  articulation  when 
the  system  is  free  from  noise. 

EFFECTS  OF  CLIPPING  ON  ARTICULATION.  Articulation  tests  that  have  been 
made  with  voice-operated  relays  give  an  indication  of  the  importance  to  articulation  of 


Initial  Consonant  Articulation  h 
M  row-^oiQivjcoioc 
o  o  ooooooo_oc 

Is* 

^3, 

^v 

X 

rx 

X 

X 

\ 

x 

\ 

Vo 

\ 

0               0.02            0.04           0.06           O.C 
Interval  of  Time  Cljpped  from  Initia 

38           0.10           0.12 
Consonant  in  Seconds 

0.1 

FIG.  8.      Effect  of  Clipping  an  Element  of  Time  from  Speech  Sounds 

initial  portions  of  the  duration  intervals  of  speech  sounds.  In  the  tests,  syllables  of  the 
consonant-vowel-consonant  type  were  spoken  at  intervals  of  about  3  sec.  A  circuit 
having  a  relay  adjusted  so  as  to  break  contact  almost  simultaneously  with  the  beginning 


ARTICULATION  TESTS 


12-35 


90 

n 

80 

/ 

~ 

i 

fa 

70 

w 

S 

t, 

f 

s 

s, 

5  60 

/ 

* 

N, 

J 

^ 

°  50 

/ 

s 

N, 

S  40 

^ 

> 

t 

30 

/ 

/ 

20 

J 

f 

10 

n 

Frequency  Multiplication  Factor 

FIG.  9.    Effect  on  Articulation  of  Multiplying  Frequencies 
by  a  Constant  Factor 


of  a  syllable  was  used.  The  contacts 
of  a  second  relay  formed  a  short  cir- 
cuit across  the  receiver.  The  oper- 
ation of  the  first  relay  caused  the 
second  relay  to  break  contact  after 
an  interval  of  time  depending  on  the 
time  constants  of  the  relay  circuit 
alone.  The  time  taken  for  the  sec- 
ond relay  to  operate  represents  the 
time  clipped  from  the  initial  con- 
sonants of  the  syllables.  Figure  8 
shows  the  initial  consonant  articula- 
tion plotted  against  the  operating 
time  of  the  second  relay.  The  data 
indicate  that  equal  elements  of  time 
in  the  duration  intervals  of  the 
sounds  were  of  equal  importance  to 
the  average  articulation.  For  ex- 
ample, the  average  duration  time  of 
the  initial  consonants  in  these  tests 
was  about  0.16  sec.  When  a  time  interval  of  0.08  sec  was  clipped  from  the  sounds,  the 
articulation  was  decreased  by  a  factor  of  about  50  per  cent. 

EFFECTS  OF  FREQUENCY  SHIFT  ON  ARTICULATION.    One  type  of  frequency 

shift,  namely,  the  multiplication 
of  frequencies  by  a  constant  fac- 
tor, occurs  when  the  speed  of  a 
sound  film  or  phonograph  turn- 
table during  reproduction  is  dif- 
ferent from  the  speed  used  during 
recording.  The  effect  of  this  type 
of  distortion  on  articulation  is 
shown  in  Fig.  9.  Another  type, 
the  addition  or  subtraction  of  a 
constant  number  of  cycles  to  all 
frequencies,  occurs  when  the  fre- 
quency of  the  carrier  at  the  mod- 
ulating end  differs  from  that  at 
the  demodulating  end  of  a  carrier 

Frequency  Shift       "  system.    The  effect  of  this  type 

Fia.  10.     Effect  of  Frequency  Shift  on  Articulation  of  shift  is  shown  in  Fig.  10.     In 

general,  the  decrease  in  articula- 
tion is  greater  when  the  frequencies  are  shifted  to  lower  values  than  when  they  are  shifted 
toward  higher  values.  In  the  first  type  of  distortion  the  duration  of  the  sounds  is 
changed,  whereas  in  the  second  type  the  duration  is  unchanged.  In  the  first  type  the  har- 
monic relationship  is  maintained,  whereas  in  the 
second  type  this  relationship  is  not  preserved.  In 
order  to  interpret  speech  sounds  perfectly  it  seems 
to  be  necessary  to  preserve  both  these  properties. 

EFFECTS  OF  PHASE  DISTORTION  ON  AR- 
TICULATION. The  aspect  of  phase  distortion  that 
has  received  most  attention  is  the  so-called  "delay 
distortion"  which  arises  when  the  phase  characteris- 
tic of  a  transmission  system  is  not  proportional  to 


1UU 

90 
80 
70 

|60 

•550 
.« 
S40 

30 
20 

10 

°3 

/ 

""•s 

\, 

/ 

x| 

\ 

^ 

S* 

"V, 

^v 

-*" 

^"x, 

^s. 

•^ 

I 

3ow 

iwar 

d 

Upv 

fard 

00      200        100          0          100       200       3OO       400        5C 

frequency  but  is  of  the  type  shown  by  the  curved    £ 
line  in  Fig.  11.    This  type  is  of  interest  because  it    •= 


Fig.  __- 

occurs  in  loaded  lines  and  in  low-pass  niters.  The 
delay  distortion  (see  Section  5,  article  9)  is  the  ^dif- 
ference between  the  slope  of  the  phase  characteristic 
at  any  frequency  /  and  the  minimum  slope,  and  is 
usually  expressed  in  the  form 


(*J\  -  (*£} 

\du/f        VSoj/ 


where  £  is  the  phase  shift  in  radians  and  w  =  2irf.      ~2 

The  approximate  effect  of  this  type  of  distortion  is    FIG.  11.    A  Curved  Phase  Characteristic 


12-36 


ACOUSTICS 


Non-Delayed  Band  0-3000 
Delayed  Band  3000-4500 


100 


E—  £L 

0                     .04 

.08                  .12                   .16                  .2 

Non -Delayed  Band  0-2000 
Delayed  Band  2000-4500 


Non-Delayed  Band  O-1500 
Delayed  Band  1500-4500 


to  delay  the  frequency  com- 
ponents near  /,  with  respect 
to  those  components  in  the 
range  of  minimum  slope,  by 
the  amount  of  the  delay  dis- 
tortion. The  effects  on  ar- 
ticulation of  delaying  various 
portions  of  the  speech  fre- 
quency range  are  shown  in 
Fig.  12.  In  these  tests,  a 
nominal  undistorted  speech 
frequency  range  of  0  to  4500 
cycles  was  divided  into  two 
parts  by  means  of  niters  and 
each  part  transmitted  through 
a  different  channel.  After 
transmission  the  two  parts 
were  recombined.  The  phase 
characteristic  of  each  channel 
approximated  a  straight  line 
over  the  greater  part  of  the 
frequency  range.  The  slope 
of  the  characteristic  of  one 
channel  could  be  increased  by 
various  amounts  over  that  of 
the  other  channel.  One  chan- 
nel thus  introduced  a  definite 
time  delay,  in  the  sense  used 
here,  with  respect  to  the  other 
channel,  i.e.,  a  delay  given  by 
the  difference  in  the  slopes  of 
their  phase  characteristics. 
The  articulation  values  de- 
crease with  increasing  delay 
and  approach  the  articulation 
of  the  more  intelligible  band 
which  was  also  the  least- 
delayed  band. 

To  determine  the  effects  on 
articulation  of  delay  distor- 
tion which  varied  continu- 
ously with  frequency,  a  sys- 
tem having  the  delay-distor- 
tion characteristics  shown  in  Fig.  13  was  studied.  These  characteristics  were  obtained  from 
network  sections  of  the  all-pass  type.  By  using  different  numbers  of  sections,  different 
amounts  of  delay  distortion  could  be  obtained.  The  attenuation  characteristics  of  the 
networks  were  equalized  to 
2500  cycles,  and  a  2400- 
cycle  low-pass  filter  having 
negligible  phase  distortion 
was  associated  with  the  net- 
works. Figure  14  shows  the 
effect  of  the  delay  distortion 
on  articulation. 

To  study  the  effects  of  de- 
lay distortion  at  higher  fre- 
quencies, articulation  tests 
were  made  on  a  system  con- 
taining first  one,  and  then 
twenty-five,  5000-cycle  low- 
pass  filters  in  series.  In  both 
cases,  the  attenuation  was 
equalized  to  5000  cycles.  "  o  400  800  1,200  i,eoo 2,000  2,400  2^800 

The   delay-distortion    char-  Frequency  In  Cycles  per  Second 

a^fceristics  and  the  results  of  Fro.  13.    Delay  Distortion  for  an  All-pas*  Network 


Non-Delayed  Band  750-4900 
Delayed  Band  0-750 


LUU  r  

H 

<j 

L 

.L 

Art! 
Non-C 

:ulat1o 
delayed 

I  Of/ 

Band 

0                    .04                   .08                   .12                   .16                 .2 
Time  of  Delay  In  Seconds 
FIG.  12.     Effects  of  Delay  on  Articulation 

ARTICULATION  TESTS 


12-37 


the  tests  are  shown  in  Fig.  15.  Because  the  component  frequencies  near  the  cutoff  are 
delayed  by  phase  distortion  of  this  type,  they  fail  to  contribute  their  normal  amounts  to 
the  total  articulation  carried  by  the  unattemiated  frequency  band.  Thus  the  distortion 
has  the  effect  of  reducing  the  transmitted  frequency  range,  'so  that  the  effective  trans- 
mitting range  depends  upon  both  the  phase  and  the  attenuation  characteristics. 


Sound  Articulation  M 

00  00  <O  U>  O 

o  01  o  01  o 

—    — 

-£-. 

~"     • 

•        II. 

sfe. 

o 

3                4                8               12               16              20               24               21 
Number  of  Sections 

FIG.  14.    Articulation  vs.  Delay  Distortion 

AUDIBLE  EFFECTS  OF  PHASE  DISTORTION.  Quite  aside  from  the  effects  on 
articulation,  when  speech  from  a  system  having  phase  distortion  is  compared  with  that 
from  a  system  of  negligible  distortion,  it  is  noticed  that  the  distorted  speech  is  axicom- 
panied  by  certain  audible  effects  which  appear  to  be  extraneous  to  the  speech  and  transient 
in  character.  These  effects,  which  are  due  to  components  of  different  frequencies  arriving 
at  different  times,  are  often  termed  "birdies"  or  **tweets,"  for  delay  distortion  of  the  type 
shown  above.  Their  noticeableness  depends  upon  the  amount  of  delay  distortion  and 
the  frequency  range  in  which  it  occurs.  For  speech,  it  was  found  that  one  section  of  the 
all-pass  network,  when  associated  with  a  2400-cycle  low-pass  filter,  had  sufficiently  small 
delay  distortion  so  as  to  be  just  noticeable.  This  determination  was  made  by  alternately 
listening  to  speech  from  the  system  under  two  conditions:  one,  the  filter  alone;  two,  the 


0.014 


CL012 


0.008 


0.006 


0.002 


1         I         I         < jOrte  Filter  ; 
Observed  Sound  Articulation       98.4 


5  Fitters  f  25  Fffiets 
97.8 


Ll5  Ftters 


10  Flters 


1,000 


5,OOO 


2,000         3,000        4,000 
Frequency  in  Cycles  per  Second 
FIG.  15.     Articulation  and  Delay  Distortion  for  5000-cyde  Lew-pass  Filters 

filter  with  the  aft-pass  network.  Judgments  of  which  condition  contained  the  network 
were  correct  about  50  per  cent  of  the  time  and  wrong  about  50  per  cent  of  the  time  for 
one  section  of  the  network.  The  total  delay  distortion  at  the  cutoff  frequency  in  this 
case,  i.e.,  that  due  to  the  filter  plus  that  due  to  one  section  of  the  network,  was  about 
0.006  see.  When  three  seefcicms  were  used  the  distortion  was  easily  noticed. 


12-38 


ACOUSTICS 


Similar  tests  with  the  5000-cycle  low-pass  filters  indicated  that  some  number  between 
5  and  10  filters  in  tandem  would  cause  just-noticeable  distortion  and  that  the  Distortion 
was  clearly  noticeable  for  20  filters  in  tandem.  The  amount  of  delay  distortion  at  the 
cutoff  frequency  for  5  filters  is  about  0.003  sec,  and  for  10  niters  about  0.006  sec 

The  above  figures  depend  somewhat  upon  the  attenuation  characteristic  as  the  cutoff 
frequency  is  approached,  small  amounts  of  attenuation  reducing  the  noticeability  of  the 
effects.  The  figures  also  vary  somewhat  with  individuals,  depending  upon  their  experi- 
ence and  hearing  characteristics. 

Tests  on  piano  reproduction  when  single  notes  were  struck  or  when  a  passage  of  music 
was  played  indicated  that  the  distortion  caused  by  25  of  the  5000-cycle  low-pass  filters  in 
tandem  was  not  noticeable.  As  in  speech,  it  would  be  expected  that  the  noticeableness 
of  the  distortion  would  depend  upon  the  frequency  range  in  which  it  occurs.  In  general, 
the  effects  of  delay  distortion  on  music  are  very  much  less  noticeable  than  on  speech, 
which  is  probably  due  in  part  to  the  more  sustained  character  of  music. 

EFFECTS  OF  ROOM  REVERBERATION  ON  ARTICULATION.  To  determine  the 
effects  of  room  reverberation  on  articulation,  the  microphone  of  a  system  substantially 


100 


100 


90 


100 


90 


a  I  Cons«.  Art- 


Stop 


468 


468 


468  24  2468 

0.1  1.0  10  0.1  1.0 

Distance  from  Transmitter- Ft. 
FIG.  16.     Effect  of  Reverberation  on  Articulation 

free  of  distortion  was  placed  in  a  room  having  variable  reverberation  conditions.  The 
room  dimensions  were  20  ft  by  30  ft  by  15  ft.  The  reverberation  time  was  practically 
uniform  over  the  important  speech  frequency  range.  The  articulation  tests  were  made 
under  three  conditions  of  reverberation,  corresponding  to  the  reverberation  times  1.2,  2.2, 
and  4.0  sec,  and  for  the  following  four  distances  between  the  diaphragm  of  the  microphone 
and  the  lips  of  the  speaker,  0.12  ft,  0.66  ft,  3.0  ft,  and  15  ft.  In  aU  the  tests,  the  speaker 
faced  toward  the  microphone.  Observers  listened  in  another  room,  by  means  of  head 
receivers,  to  the  speech  picked  up  by  the  microphone  in  the  reverberant  room.  A  volume 
indicator  was  used  to  measure  the  level  of  the  speech  from  the  microphone.  The  instru- 
ment measures,  substantially,  the  phonetic  power  of  the  vowel  sounds.  The  gain  of  the 
system  between  microphone  and  receiver  was  set  so  that  the  level  of  the  speech  received 
by  the  observers,  for  the  close. talking  condition,  was  74  db  above  threshold.  The  gain 
of  the  system  was  then  kept  constant  for  all  tests.  The  loss  in  speech  level  vs.  distance 
between  microphone  and  speaker,  and  the  articulation  results,  are  shown  in  Fig.  16.  The 
speech  level  decreased  inversely  as  the  square  of  the  distance  for  distances  up  to  1  ft  from 
the  transmitter,  but  at  the  15-ft  distance  the  level  was  some  12  db  greater  than  the  level 
obtained  by  assuming  the  inverse  square  relation.  The  solid  curves  designated  as  "sys." 


ACOUSTIC  PROPERTIES   OF  ROOMS  12-39 

show  articulation  vs.  distance  between  speaker  and  microphone,  for  the  system  alone. 
They  are  graphs  of  the  data  of  Figs.  3  and  4,  where  level  above  threshold  has  been  con- 
verted into  distance  between  speaker  and  microphone.  This  was  done  by  means  of  the 
solid  curve  showing  loss  in  speech  level  vs.  distance,  and  taking  the  level  for  zero  loss  as 
£  t  6  threshold-  This  level  is,  of  course,  some  20  db  less  than  the  level  the  observer 
would  have  received  if  the  speaker  had  spoken  directly  into  his  ear  from  a  distance  of  0. 12  ft. 
A  comparison  of  the  curves  indicates  that  the  various  sound  groups  were  not  uniformly 
attenuated  by  increasing  the  distance  between  speaker  and  microphone.  For  a  given 
distance,  the  effect  of  increasing  the  reverberation  time  was  always  in  the  direction  of 
decreasing  the  articulation,  even  though  the  received  speech  levels  increased  as  the  rever- 
veration  increased.  For  distances  up  to  15  ft,  at  least,  the  articulation  was  appreciably 
less  for  a  reverberant  room  than  it  would  be  for  one  with  perfectly  absorbing  walls. 

15.  AUDITORY  PERSPECTIVE 

Part  of  the  emotional  and  esthetic  appeal  of  sounds,  when  listened  to  directly  with  both 
ears,  may  be  ascribed  to  the  appreciation  of  their  spatial  character.  When  this  property 
of  sound  waves  is  preserved  in  reproduction,  the  sounds  are  said  to  be  reproduced  in  true 
auditory  perspective.  Ideally,  there  are  two  ways  of  accomplishing  such  reproduction. 
One  is  binaural  reproduction,  which  aims  to  reproduce  in  the  distant  listener's  ears,  by 
means  of  head  receivers,  exact  copies  of  the  sound  waves  that  would  exist  in  his  ears  if  he 
were  listening  directly.  In  this  case,  a  complete  system  consisting  of  microphone,  line, 
and  receiver  is  used  for  each  ear.  The  other  method  uses  loudspeakers  and  aims  to  repro- 
duce in  the  distant  hall  an  exact  copy  of  the  pattern  of  sound  vibration  that  exists  in 
the  original  hall.  In  the  limit,  an  infinite  number  of  microphones  and  loudspeakers  of 
infinitesimal  dimensions  would  be  needed.  In  a  symposium  on  reproduction  in  auditory 
perspective  the  basic  requirements  for  the  production  of  orchestral  music  were  discussed 
and  a  practical  system  designed  to  achieve  auditory  perspective  was  described  (Bell 
Sys.  Tech.  J.,  April  1934). 

BIBLIOGRAPHY 

Auditory  Perspective,  a  collection  of  six  papers,  BeZZ  Sys.  Tech.  J".,  April  1934,  p.  239. 

Fletcher,  An  Acoustic  Illusion  TelephonicalLy  Achieved,  BetL  Lab.  Rec.,  June  1933,  p.  286. 

Gait,  R.  H.f  Noise  Out-of-doors,  J.  Acous  Soc.  Am.,  July  1930,  p.  30. 

Snow,  W.  B.,  Audible  Frequency  Ranges  of  Music,  Speech,  and  Noise,  J.  Acous.  Soc.  Am.,  July  1931, 

p.  155. 
Steinberg,  J.  C.,  Effects  of  Distortion  on  the  Recognition  of  Speech  Sounds,  J.  Aco-u*.  Soc.  Am.,  October 

1929,  p.  121. 

Steinberg,  J.  C.,  Effects  of  Phase  Distortion  on  Telephone  Quality,  Bdl  Sys.  Tech.  J.t  July  1930,  p.  550. 
Steinberg  and  Snow,  Auditory  Perspective  and  the  Physical  Factors  Affecting  It,  BeU,  Sys.  Tech.  J., 

April  1934,  p.  245. 
Tucker,  R.  S.,  Noise  in  Buildings,  J.  Acous.  Soc.  Am.,  July  1930,  p.  59. 

ACOUSTIC  PROPERTIES  OF  ROOMS 

By  Vern  O.  Knudsen 

The  acoustic  quality  of  nearly  all  communication  systems,  as  radio,  television,  teleph- 
ony, motion  pictures,  phonograph  recording,  hearing  aids,  and  public-address  and  other 
sound-amplifying  or  -reproducing  systems,  is  largely  influenced  by  the  acoustic  properties 
of  the  rooms  in  which  the  sound  to  be  transmitted  or  recorded  is  generated,  and  in  which 
this  sound  is  reproduced. 

The  following  definitions  pertinent  to  the  acoustic  properties  of  rooms  have  been  selected 
from  American  Standard  Acoustical  Terminology — Z24.1-1942-(see  J.  Acous.  Soc.  Am., 
Vol.  14,  84-110  [1942]  for  these  and  other  pertinent  definitions). 

Effective  Sound  Press-ore  (P).  The  effective  sound  pressure  at  a  point  is  the  root 
mean  square  value  of  the  instantaneous  sound  pressure  over  a  complete  cycle,  at  that 
point.  The  unit  is  the  dyne  per  square  centimeter. 

Pressure  Level.*  The  pressure  level,  in  decibels,  of  a  sound  is  20  times  the  logarithm 
to  the  base  10  of  the  ratio  of  the  pressure  P  of  this  sound  to  the  reference  pressure  PQ.  Un- 
less otherwise  specified,  the  reference  pressure  is  understood  to  be  0.0002  dyne  per  sq  cm. 

*  In  discussing  sound  measurements  made  with  pressure  or  velocity  microphones,  especially _  in 


and  the  pressure  or  velocity  is  generally  unknown. 


12-40  AOOWSTICS 

Velocity  Level.*  The  velocity  level,  in  decibels,  -of  a  sound  is  20  times  the  logarithm 
to  the  base  10  of  the  ratio  of  the  particle  velocity  of  the  sound  to  the  reference  particle 
velocity.  Unless  otherwise  specified  the  reference  particle  velocity  is  understood  to  be 
5  X  10"6  cm  per  sec  effective  value. 

Sound  Energy  Density  (£).  Sound  energy  density  is  the  sound  energy  per  unit  volume. 
The  unit  is  the  erg  per  cubic  centimeter. 

Sound  Intensity  *  (/).  The  sound  intensity  of  a  sound  field  in  a  specified  direction 
at  a  point  is  the  sound  energy  transmitted  per  unit  of  time  'in  the  specified  direction  through 
a  unit  area  normal  to  this  direction  at  the  point.  The  unit  is  the  erg  per  second  per  square 
centimeter,  but  sound  intensity  may  also  be  expressed  in  watts  per  square  centimeter. 

Intensity  Level  *  (/£>.  The  intensity  level,  in  decibels,  of  a  sound  is  10  times  the 
logarithm  to  the  base  10  of  the  ratio  of  the  intensity  /  of  this  sound  to  the  reference  in- 
tensity /o.  Unless  otherwise  specified  the  reference  intensity  J0  shall  be  10  16  watt  per 
sq  cm. 

Echo.  An  echo  is  a  wave  which  has  been  reflected  or  otherwise  returned  with  suffi- 
cient magnitude  and  delay  to  be  perceived  in  some  manner  as  a  wave  distinct  from  that 
directly  transmitted. 

Multiple  Echo.  A  multiple  echo  is  a  succession  of  separately  distinguishable  echoes 
from  a  single  source. 

Fhrtter  Echo.  A  flutter  echo  is  a  rapid  succession  of  reflected  pulses  resulting  from  a 
single  initial  pulse.  If  the  flutter  echo  is  periodic  and  if  the  frequency  is  in  the  audible 
range  it  is  called  a  musical  echo. 

Noise.    Noise  is  any  undesired  sound. 

Acoustic  Reflectivity.  The  acoustic  reflectivity  of  a  surface  not  a  generator  is  the 
ratio  of  the  rate  of  flow  of  sound  energy  reflected  from  the  surface,  on  the  side  of  incidence, 
to  the  incident  rate  of  flow.  Unless  otherwise  specified,  an  possible  directions  of  incident 
flow  are  assumed  to  be  equally  probable.  Also,  unless  otherwise  stated,  the  values  given 
apply  to  a  portion  of  an  infinite  surface,  thus  eliminating  edge  effects. 

Acoustic  Absorptivity.  The  acoustic  absorptivity  of  a  surface  is  equal  to  1  minus  the 
reflectivity  of  that  surface.  (Sometimes  called  Sound-absorption  Coefficient.} 

Sabin.  The  sabin  is  a  unit  of  equivalent  absorption;  it  is  equal  to  the  equivalent  ab- 
sorption of  1  sq  ft  of  a  surface  of  unity  absorptivity,  i.e.,  of  1  sq  ft  of  surface  which  ab- 
sorbs all  incident  sound  energy. 

Acoustic  Transmittivity.  The  acoustic  transmittivity  of  an  interface  or  septum  is  the 
ratio  of  the  rate  of  flow  of  transmitted  sound  energy  to  the  rate  of  incident  flow.  Unless 
otherwise  specified,  all  directions  of  incident  flow  are  assumed  to  be  equally  probable. 

Reverberation.     Reverberation  is  the  persistence  of  sound,  due  to  repeated  reflections. 

Rate  of  Decay  (of  Sound  Energy  Density).  The  rate  of  decay  of  sound  energy  density 
is  the  time  rate  at  which  the  sound  energy  density  is  decreasing  at  a  given  point  and  at 
a  given  time.  The  practical  unit  is  the  decibel  per  second. 

Reverberation  Time  (T),  The  reverberation  time  for  a  given  frequency  is  the  tune 
required  for  the  average  sound  energy  density,  initially  in  a  steady  state,  to  decrease, 
after  the  source  is  stopped,  to  one-millionth  of  its  initial  value.  The  unit  is  the  second. 

Mean  Free  Path.  The  mean  free  path  for  sound  waves  in  an  in  closure  is  the  average 
distance  sound  travels  in  the  inclosure  between  successive  reflections, 

16.  REQUIREMENTS  FOR  GOOD  ACOUSTICS 

In  order  that  a  room  may  have  good  acoustics  it  is  necessary  (1)  that  the  sound  be 
sufficiently  loud  in  the  room;  (2)  that  the  room  be  free  from  noise  whether  of  internal  or 
external  origin;  (3)  that  the  room  be  free  from  echoes,  "flutters,"  or  other  interfering 
reflections;  (4)  that  the  reflecting  boundaries  of  the  room  be  so  disposed  as  to  provide  a 
nearly  uniform  distribution  of  sound  energy  throughout  the  room;  (5)  that  the  room  be 
free  from  undesirable  resonance;  and  (6)  that  the  reverberation  in  the  room  be  sufficiently 
reduced  to  avoid  excessive  overlapping  or  commingling  of  successive  sounds  of  articulated 
speech  or  music,  but  that  the  room  be  sufficiently  "live"  at  all  frequencies  to  give  a  pleas- 
ing effect  to  either  speech  or  music  as  judged  by  the  average  listener. 

In  order  to  attain  these  necessary  conditions  for  good  acoustics  the  architect  and  acousti- 
cal engineer  must  assume  responsibility  relating  to  the  following:  (1)  the  selection  of  the 

*Ia  discussing  sound  measurements  made  with  pressure  or  velocity  microphones,  especially  in 
imeJosures  involving  normal  modes  of  vibration  or  in  sound  fields  containing  standing  waves,  caution 
must  be  observed  in  using  the  terms  "intensity"  and  "intensity  leveL"  Under  such  conditions  the 
teraaa  *  pressure  level"  or  "velocity  level"  are  preferable  since  the  relaiaonsnip  between  the  intensity 
and  the  pressure  or  velocity  is  generally  unknown. 


GEOMETRIC  AND  WAVE  ACOUSTICS  12-41 

site;  (2)  the  making  of  a  noise  survey  in  the  proximity  of  the  proposed  site;  (3)  the  selec- 
tion ol  a  general  type  of  wall  and  ceiling  construction  which  will  insulate  the  building 
adequately  against  external  noise  and  vibration;  (4)  the  selection  and  arrangement  of 
rooms  which  require  acoustical  designing;  (5)  the  design  of  the  rough  sketches  for  all 
speech  rooms,  music  rooms,  recording  or  broadcasting  rooms,  based  upon  the  requirements 
for  the  proper  distribution  of  direct  and  of  reflected  sound;  (6)  the  application  of  appro- 
priate formulas  and  principles  to  the  detailed  design  of  shape,  sound  insulation,  and  sound 
absorption  for^all  rooms  which  require  acoustical  designing;  (7)  the  selection  of  materials 
which  will  satisfy  the  acoustical,  structural,  decorative,  and  economic  requirements;  (8) 
the  supervision  of  all  aspects  of  construction  which  will  affect  the  outcome  in  acoustics, 
and^  especially  the  making  of  tests  on  such  materials  as  acoustical  plaster;  and  (9)  the 
testing  of  the  completed  building  with  regard  to  the  distribution  of  sound;  freedom  from 
echoes,  sound  foci,  or  interfering  reflections;  the  optimal  conditions  of  reverberation;  and 
the  adequacy  of  sound  insulation.  In  general,  the  acoustic  problem  consists  of  the  ade- 
quate reduction  of  noise  and  vibration,  and  the  designing  of  interiors  in  which  the  voice  or 
instrumentation  is  heard  or  recorded  most  satisfactorily.  All  the  factors  mentioned  in  this 
section  relate  to  the  problem  of  room  acoustics.  One  of  the  most  basic  of  these  factors  is 
the  problem  of  the  growth  and  decay  of  sound  in  a  room  or,  more  strictly,  the  transient 
and  steady-state  behavior  of  sound  in  that  room.  Before  considering  this  problem,  it  will 
be  helpful  to  describe  briefly  two  possible  approaches  to  this  and  related  problems:  one, 
geometric  acoustics,  and  the  other,  wave  acoustics, 

17.  GEOMETRIC  AND  WAVE  ACOUSTICS 

Geometric  or  "ray"  acoustics,  as  applied  to  the  acoustics  of  rooms,  assumes  that  sound 
travels  in  rays,  that  its  frequency  remains  unchanged  during  the  transient  state  as  well 
as  the  steady  state,  that  the  rays  are  reflected,  with  partial  absorption  and  transmission, 
at  each  encounter  with  the  boundaries  of  the  room,  and  that  after  a  number  of  successive 
reflections  the  sound  in  the  room  becomes  diffuse  (all  directions  of  propagation  being 
equally  probable)  and  of  uniform  energy  density  throughout  the  room.  Obviously,  this 
oversimplifies  the  actual  behavior  of  sound  waves  in  the  room,  especially  when  the  wave- 
lengths of  the  sound,  as  is  often  the  case,  are  not  small  compared  with  the  dimensions  of 
the  room;  it  neglects  entirely  such  important  properties  as  the  normal  modes  of  vibration 
of  the  room,  interference,  and  diffraction.  In  spite  of  these  shortcomings,  it  leads  to  many 
principles  and  formulas  by  means  of  which  the  acoustical  engineer  or  architect  can  design 
auditoriums  with  satisfactory  acoustics. 

Wave  acoustics  deals  with  sound  as  waves;  it  offers  the  only  means  of  dealing  rigorously 
with  wave  phenomena  in  bounded  spaces,  with  interference,  diffraction,  normal  modes  of 
vibration,  and  with  the  influence  of  localized  areas  of  absorptive  material  and  of  irregular- 
ities of  the  contours  of  the  boundaries  of  the  room  on  both  the  transient  and  steady  states 
of  the  sound  in  the  room.  The  difficulties  in  applying  wave  acoustics  to  rooms  are  so 
great  that  very  little  progress  has  been  made  until  recently,  and  even  now  much  remains 
to  be  done  before  it  can  be  used  effectively  by  the  average  architect  or  acoustical  engineer. 

Historically,  the  study  of  the  acoustic  properties  of  rooms  began  and  has  continued, 
very  largely,  on  the  assumption  that  geometric  acoustics  is  adequate  to  cope,  at  least 
approximately,  with  the  transient  and  steady-state  behavior  of  sound  waves  in  most  rooms. 
Obviously,  the  methods  of  geometric  acoustics  furnished  satisfactory  approximations  in 
rooms  in  which  the  ratio  of  the  room  dimensions  to  the  wavelength  of  the  sound  is  large 
and  in  which  the  sound  distribution  is  thoroughly  diffuse  (the  "ergodic"  state),  but  in 
most  rooms,  and  especially  for  sounds  of  low  frequency,  the  more  rigorous  methods  of 
wave  acoustics,  recently  developed  by  Morse  and  Bolt,  should  be  used  so  far  as  this  is 
possible.  (See  Morse  and  Bolt,  Sound  Waves  in  Rooms,  Remews  «/  Modern,  Posies, 
April  1944).  These  methods  of  wave  acoustics  already  have  explained  many  apparent 
anomalies  in  room  acoustics,  especially  those  relating  to  the  absorptive  properties  of 
acoustic  materials  as  used  in  different  rooms.  No  one  who  is  not  familiar  with  these 
methods  should  undertake  the  design  of  a  room  in  which  good  acoustics  is  a  prime  require- 
ment. Unfortunately,  the  results  of  these  studies  are  not  yet  sufficiently  simplified  to  be 
used  by  the  average  engineer  as  the  basis  for  making  routine  analyses  of  the  acoustic 
properties  of  rooms.  Fortunately,  on  the  other  hand,  the  approximate  and  much  simpler 
methods  of  geometric  acoustics,  when  used  with  an  understanding  of  the  consequences 
and  modifications  which  wave  acoustics  entails,  will  be  found  to  be  satisfactory  for  making 
the  routine  calculations  which  govern  the  acoustic  properties  of  most  rooms,  such  as  offices, 
restaurants,  classrooms,  and  residential  rooms.  For  the  acoustic  design  of  broadcasting 
and  motion-picture  studios,  auditoriums,  theaters,  churches,  music  rooms,  courtrooms, 


12-42  ACOUSTICS 

lecture  rooms,  and  all  other  rooms  in  which  high-quality  speech  and  music  are  required, 
full  use  should  be  made  of  the  methods  of  wave  acoustics,  as  far  as  they  are  applicable. 

For  the  present,  the  methods  of  geometric  acoustics  continue  to  be  used  by  most  engi- 
neers for  designing  or  correcting  the  acoustic  properties  of  rooms,  and  their  methods 
probably  will  continue  to  be  used  for  possibly  another  five  to  ten  years,  but  they  should 
be  used  only  in  the  light  of  existing  and  advancing  knowledge  of  wave  acoustics.  In  the 
following  treatment,  based  largely  on  geometric  acoustics,  the  practical  modifications 
which  wave  acoustics  imposes  or  suggests  will  be  considered,  and  those  who  use  this  hand- 
book as  a  guide  for  planning  for  good  acoustics  should  give  similar  consideration  to  the 
relevance  of  wave  acoustics  in  modifying  the  calculations  and  conclusions  based  on  geo- 
metric acoustics. 

18.  GROWTH  AND  DECAY  OF  SOUND  IN  ROOMS- 
GENERAL  CONSIDERATIONS 

The  approximate  theories  in  use  today  are  based  upon  the  assumptions  that  sound, 
originating  at  some  point  in  a  room,  propagates  rays  of  vibratory  energy  with  a  speed  of 
about  1125  ft  per  sec,  uniformly  in  all  directions;  that  these  rays  are  partially  reflected 
by  the  boundaries  of  the  room;  and  that  even  after  the  source  of  sound  is  stopped  these 
rays  persist  with  their  original  frequency  but  become  feebler  after  each  reflection  until 
ultimately  they  become  inaudible.  In  these  approximate  theories  it  is  assumed  that  the 
sound  energy  persists  in  rays  or  bundles;  that,  during  the  decay,  the  sound  energy  in  the 
room  remains  constant  in  these  rays  or  bundles  for  a  short  interval  of  time,  equal  to  the 
time  required  for  a  ray  to  travel  the  average  distance  between  successive  reflections,  the 
mean  free  path,  and  that  then  the  total  sound  energy  in  the  room  suddenly  drops  a  certain 
amount  determined  by  the  "average"  (usually  the  weighted  arithmetical  mean)  acoustic 
absorptivity  of  the  boundaries  of  the  room;  and  that  this  process  of  absorption  by  discrete 
steps  continues  until  all  the  sound  energy  is  converted  into  heat. 

Although  most  of  the  absorption  takes  place  at  the  boundaries  at  low  frequencies,  the 
absorption  in  the  air  at  frequencies  above  5000  cycles  may  be  greater  than  the  absorption 
at  the  boundaries.  If  the  source  continues  to  generate  sound  at  a  constant  rate,  a  condi- 
tion of  equilibrium  will  be  reached  in  which  the  rate  of  supply  of  sound  energy  to  the  room 
is  just  equal  to  the  rate  of  absorption  by  the  air  and  the  boundaries.  If  the  source  is  then 
stopped  the  sound  in  the  room  will  die  away  at  a  rate  equal  to  the  rate  of  absorption,  which 
is  determined  principally  by  the  size,  the  shape,  and  the  boundaries  of  the  room.  Although 
this  decay  is  strictly  made  up  of  the  free,  damped,  normal  modes  of  vibration  of  a  three- 
dimensional  continuum,  the  decay  is  approximately  represented  by  the  simplification  de- 
scribed above,  provided  that  the  absorptive  material  is  distributed  throughout  the  bound- 
aries of  the  room  and  especially  that  it  is  not  concentrated  on  one  or  two  walls  of  the  room. 
(Walls,  as  here  used,  refers  also  to  the  floor  and  ceiling.) 

According  to  this  simplification,  the  time  required  for  the  intensity  of  the  sound  to  be 
reduced  a  specified  amount  will  depend  upon  (1)  the  number  of  reflections  which  occur 
per  unit  time,  and  (2)  the  amount  of  sound  energy  which  is  absorbed  at  each  reflection. 
If  the  room  is  a  large  one  there  will  be  only  a  few  reflections  per  second;  and  in  addition, 
if  but  a  little  sound  energy  is  absorbed  at  each  reflection,  it  will  require  a  relatively  long 
time  for  the  intensity  of  ordinary  sound  to  be  reduced  to  the  threshold  of  audibility. 
Such  a  room  will  be  excessively  reverberant.  On  the  other  hand,  if  the  room  is  small  and 
the  boundaries  highly  absorptive,  the  room  will  be  free  from  reverberation.  Since  the 
average  intensity  of  speech  or  music  in  a  room  is  of  the  order  of  one  million  times  the  in- 
tensity which  is  just  barely  audible,  and  since  the  hard,  rigid  boundaries  may  reflect  as 
much  as  98  per  cent  of  the  incident  sound  energy,  it  is  apparent  that  an  appreciable  time, 
amounting  to  several  seconds  in  many  instances,  is  necessary  for  the  sounds  of  speech  or 
music  to  be  reduced  to  inaudibility. 

Thus,  consider  a  room  having  a  mean  free  path  of  51  ft.  Since  the  velocity  of  sound  at 
room  temperature  (20  deg  cent)  is  approximately  1122  ft  per  sec,  there  will  be  in  this  room 
just  22  reflections  each  second.  Hence,  if  the  initial  sound  in  this  room  has  an  intensity 
of  one  million  threshold  units,  and  if  98  per  cent  of  the  initial  sound  energy  is  reflected  at 
each  encounter  with  the  boundaries,  to  reduce  the  sound  energy  to  one-millionth  of  its 
initial  amount  would  require  n  successive  reflections,  where  n  is  given  by  0.98n  =  0.000001. 
Solving,  n  is  684;  that  is,  it  requires  684  successive  reflections  in  this  room  for  the  sound 
energy  to  die  away  to  inaudibility.  Since  22  reflections  occur  each  second  in  this  room, 
the  time  required  for  the  sound  to  die  away  to  inaudibility  is  684  ~  22,  or  31.1  sec;  that 
is,  the  time  of  reverberation  in  this  room  is  31.1  sec.  By  a  similar  consideration  it  can  be 
shown  that  if  this  same  room  were  completely  lined  with  a  material  that  reflects  50  per 


REVERBERATION  EQUATIONS 


12-43 


cent  of  the  incident  sound  energy  the  total  number  of  reflections  would  be  reduced  to  19.9, 
and  the  time  of  reverberation  would  be  of  the  order  of  0.9  sec.  These  simple  considera- 
tions neglect  the  absorption  of  sound  in  the  air,  which  at  high  frequencies  win  greatly 
modify  these  calculations.  This  type  of  absorption  will  be  considered  later  in  this  section, 
ine  tormulas  to  which  such  approximate  theories  lead  are  sufficiently  valid  for  most 
practical  purposes  in  rooms  which  are  bounded  by  materials  having  the  same  coefficient 
of  absorption.  However,  in  rooms  bounded  by  materials  having  widely  different  coeffi- 
cients _of  absorption,  the  formulas  approach  validity  only  as  the  decadent  sound  in  the 
room  is  made  to  approach  a  completely  diffuse  state.  It  should  be  clearly  recognized 
therefore  that  the  formulas  which  will  now  be  presented  must  be  used  with  caution  and 
understanding,  especially  with  reference  to  the  average  coefficient  of  absorption  in  rooms 
which  are  bounded  partly  by  highly  reflective  and  partly  by  highly  absorptive  materials. 


19.  REVERBERATION  EQUATIONS 

The  early  experiments  of  W.  C.  Sabine  show  that  the  time  of  reverberation  in  a  room  is 
proportional  directly  to  the  volume  of  the  room  and  inversely  to  the  total  amount  of 
absorption  supplied  by  the  boundaries  of  the  room.  Sabine  was  able  to  determine  exper- 
imentally the  constant  of  proportionality  k  between  reverberation  time  T  and  the  volume 
V  divided  by  total  absorption  a.  Thus, 

*-? 

The  value  of  k  as  determined  experimentally  by  Sabine  for  a  large  number  of  rooms  of 
different  shapes  and  sizes,  at  normal  room  temperatures,  is  approximately  0.05  when  V 
is  in  cubic  feet  and  a  is  in  sabins  (British  units)  and  0.164  when  F  is  in  cubic  meters  and  a 
is  in  square  meters  (metric  units).  A  few  years  later,  Franklin  and  also  Jaeger  obtained 
this  same  equation  from  theoretical  considerations. 

This  equation  was  used  for  nearly  30  years  for  calculating  the  reverberation  time  of 
either  contemplated  or  finished  rooms.  Even  the  fallacious  conclusion  to  which  the  equa- 
tion leads  for  a  room  with  totally  absorptive  surfaces,  namely,  that  T  =  kV/S  instead  of 
zero  (where  S  is  the  total  surface  area  of  the  room),  was  overlooked  or  was  not  sufficiently 
disturbing  to  destroy  confidence  in  its  validity,  until  recently.  The  equation  is  satisfac- 
tory in  practice  for  frequencies  between  about  200  and  1000  cycles  in  the  large  majority  of 
rooms  in  which  the  absorptive  material  is  distributed  and  in  which  the  rate  of  decay  of 
sound  is  slow;  that  is,  the  equation  applies  to  "live  rooms,"  provided  the  frequencies  are 
high  enough  to  be  well  above  the  fundamental  resonant  frequency  of  the  room  but  not 
high  enough  to  involve  a  consideration  of  the  attenuation  in  the  medium, 

MODIFICATION  OF  THE  REVERBERATION  FORMULA.  A  more  satisfactory 
reverberation  formula  can  be  obtained  by  recognizing  that  the  decay  of  sound  takes  place 
discontimtously,  at  time  intervals  equal  to  the  time  required  for  sound  to  travel  the  mean 
free  path.  The  mean  free  path  is  given  for  rooms  of  conventional  shape  as  4V/ S.  Strictly 
it  is  dependent  upon  the  shape  of  the 
room  and  the  location  of  the  source,  'OBO 
so  that  the  equation  is  not  accurate 
for  rooms  of  peculiar  shape.  Thus,  ^ 
for  the  usual  location  of  the  source  of  t  -015 
sound  and  for  the  first  25  successive  JJ 
reflections,  the  mean  free  path  is  "| 
4.3V /S  for  a  large  church  or  cathe-  J  ~0io 
dr al  of  cruciform  shape  and  is  3.7 V/S  g 
for  a  room  with  large  horizontal  di-  s. 
mensions  and  a  low  ceiling,  the  usual  g  .005 
shape  for  large  office  and  work  ^ 
rooms. 

Allowance  must  also  be  made  for 
the  absorption  of  sound  in  the  air  of 


100 


20  40  60  80 

Peroet*  Retake  ycrnfeffi^  at-fiG0  (X 

FIG.  1.    Coefficients  of  Absorption  of  Sound  in  Air  Coxt- 

tainine  Different  Amounts  of  Water  Vapor,  for  Frequencies 

of  1500,  3000,  6000,  and  10,000  Cycles 


the  room,  which  is  of  considerable 

importance   at   the  higher   audible 

frequencies  and  especially  in  large 

rooms.    The  curves  of  Fig.  1  give 

the  absorption  coefScient  m  per  foot  for  plane  waves  in  air  at  20  deg  cent.    It  will  be  seen 

from  the  curves  in  Fig.  1  that  m  has  a  maximum  for  a  certain  concentration  of  water 

vapor,  different  for  each  frequency. 


12-44  ACOUSTICS 

Based  upon  the  inclusion  of  these  factors  and  a  mean  free  path  of  4F/3  the  time  of 
reverberation  is 

(2) 
-S)]  ' 

where  c  is  the  velocity  of  sound  and  5  is  the  arithmetic  mean  of  the  absorption  coefficients 
of  all  the  boundaries  of  the  room. 

At  room  temperature,  21  deg  cent,  c  —  1125  ft  per  sec,  so  that  for  most  working  condi- 
tions 

T  =  _  2^51  _  (3) 

AmV  -  Sbi  <l  -  a)  ^' 

in  British  units  or 


in  metric  units  (see  above).  For  frequencies  below  about  1000  cycles,  m  is  so  small  that 
the  first  term  in  the  denominator  of  eqs.  (2),  (3),  or  (4)  can  be  neglected,  that  is,  the  ab- 
sorption in  the  air  is  inappreciable;  whereas,  at  high  frequencies  (above  5000  cycles), 
this  term  may  become  larger  than  the  second  (or  surface  absorption)  term.  At  sufficiently 
high  frequencies  (above  the  audible  range),  the  second  term  will  become  negligible,  in 
which  case  the  rate  of  decay  and  consequently  the  time  of  reverberation  will  be  independent 
of  the  size  of  the  room. 

The  foregoing  reverberation  formula,  eq.  (2),  is  sufficiently  valid  for  practical  purposes 
provided  the  sound  in  the  room  is  thoroughly  diffuse  throughout  the  decay.  This  condi- 
tion is  realized  for  frequencies  above  about  250  cycles  in  all  but  very  small  rooms,  provided 
all  the  boundaries  of,  the  room  have  approximately  the  same  absorptivity,  or  provided 
suitable  rotating  paddles  or  "warble  tones"  are  used  to  "mix"  the  sound  in  the  room. 
Suitable  precautions,  such  as  those  just  mentioned,  can  be  taken  in  making  measurements 
in  an  acoustical  laboratory,  such  as  a  reverberation  chamber.  In  many  rooms  encountered 
in  practice,  the  absorptive  material  may  be  concentrated  on  a  single  surface,  as  when  a 
carpet,  upholstered  seats,  and  audience  are  all  located  on  the  floor,  and  the  other  surfaces 
in  the  room  are  highly  reflective.  In  such  rooms,  especially  if  the  opposite  walls  are  paral- 
lel and  not  too  far  apart,  the  decay  of  sound  will  not  conform  to  the  approximately  expo- 
nential decay  predicted  by  eq.  (2)  but  will  consist,  first,  of  a  rapid  rate  of  decay  while  the 
sound  is  relatively  diffuse,  and  second,  of  a  much  slower  rate  of  decay,  made  up  largely 
of  a  horizontal  now  of  sound  energy  between  the  parallel  and  highly  reflective  walls,  i.e., 
of  modes  of  vibration  which  are  directed  at  grazing  incidence  to  the  floor.  The  time  of 
reverberation  in  such  a  room  will  be  longer  than  that  calculated  by  means  of  eq.  (2), 
using  an  arithmetical  mean  for  5. 

This  is  borne  out  by  some  oscillograms  of  the  rate  of  decay  obtained  in  a  small  room, 
8  ft  by  8  ft  by  9.5  ft  (high),  with  the  floor  covered  with  a  material  having  a  rated  absorp- 
tion coefficient  of  0.60  at  512  cycles  and  with  the  walls  and  ceiling  finished  with  painted 
concrete.  The  first  part  of  the  decay  (15  to  17  db)  was  relatively  rapid,  95  to  100  db  per 
sec.  This  was  followed  by  a  much  slower  decay,  38  to  40  db  per  sec.  If  the  first  part  of 
the  decay  is  used  for  calculating  the  time  of  reverberation  T  and  the  absorptivity  of  the 
floor  material  or,  we  obtain  T  =  0.61  sec  and  a  =  0.55.  If  the  latter  part  is  used,  T  «  1.54 
sec  and  a.  =  0.22.  It  will  be  noted  that  the  first  part  of  the  decay,  while  the  sound  is 
relatively  diffuse,  yields  a  value  for  a  which  agrees  fairly  well  with  the  rated  value  of  0.60 
and  therefore  conforms  reasonably  well  with  the  requirements  of  eq.  (2),  whereas  the 
latter  part  of  the  decay,  which  is  made  up  largely  of  the  horizontal  modes  of  vibration, 
is  much  slower  than  would  be  predicted  by  eq.  (2).  This  is  an  extreme  example  of  the 
inadequacy  of  eq.  (2)-,  which  is  based  on  geometric  acoustics,  to  account  for  the  true  nature 
of  the  decay  of  sound  in  a  room  in  which  the  absorptive  material  is  concentrated  on  one 
wall  (or  floor  or  ceiling).  Wave  acoustics,  on  the  other  hand,  accounts  for  the  observed 
results  very  satisfactorily. 

Fortunately,  for  the  best  acoustical  quality  in  a  room  the  absorptive  material  should  be 
distributed  on  all  surfaces  of  a  room  so  that  the  rate  of  decay  will  be  at  least  approximately 
the  same  in  all  directions,  and  under  these  circumstances  eq.  (2)  will  yield  results  which, 
as  a  rule,  do  not  differ  more  than  10  per  cent  from  the  observed  values.  Furthermore,  in 
very  large  rooms,  as  in  theaters,  school  auditoriums,  and  churches,  there  is  very  little 
tendency  for  the  reverberation  to  persist  in  two  dimensions,  even  though  most,  of  the 
absorption  is  concentrated  on  the  floor  or  on  the  floor  and  in  the  ceiling,  (1)  because  the 
dimensions  of  the  room  are  large  compared  with  the  wavelengths  of  the  sound,  and  (2) 
beeaiase*  the  'architectural  treatment  of  large  rooms  usually  involves  structural  forms  and 
ornamentations  which  tend  to  diffuse  the  sound  during  free  decay. 


ROOM  RESONANCE  12-45 

In  such  rooms,  provided  there  are  no  curved  surfaces  giving  rise  to  concentrations  of 
sound,  the  nrst  SO  db  (or  more)  of  decay  conforms  very  closely  to  eq.  (2);  and  it  is  this 
portion  of  the  decay  that  is  pertinent  to  the  acoustic  quality  of  speech  and  music  in 
rooms.  Stated  otherwise,  the  rate  of  decay  after  the  first  30  db  of  decay  is  of  little  conse- 
quence, since  in  articulated  speech  or  music  such  residual  sounds  will  be  so  weak  as  to  be 
completely  masked  by  the  primary  (and  much  louder)  sounds  that  follow.  It  is  apparent, 
therefore,  that  eq.  (2)  is  satisfactorily  valid  for  the  practical  calculations  of  reverberations 
in  most  rooms.  In  many  small  rooms,  such  as  are  frequently  used  for  radio  broadcasting 
or  for  the  recording  of  sound,  it  is  important  that  the  sound-absorptive  materials  be  dis- 
tributed quite  uniformly  throughout  the  room  if  eq.  (2)  is  to  be  applicable.  The  true  rate 
of  decay  of  sound  in  small  rooms  is  greatly  dependent  upon  room  resonance. 

20.  ROOM  RESONANCE 

A  room  is  a  resonant  chamber,  with  resonant  properties  similar  to  those  of  a  violin 
string,  an  organ  pipe,  or  a  diaphragm  of  a  telephone  receiver,  except  that  in  general  the 
resonant  properties  of  the  room  are  much  more  complicated  than  those  of  one-  or  two- 
dimensional  systems.  Thus,  if  the  dimensions  of  a  rectangular  room  are  Ii,  Zj,  and  Zj,  tbe 
"resonant"  or  normal  frequencies  v  for  the  room  are  given  approximately  by 

-        c  \  If       If 
"  =  2U?  +  ^  + 

where  c  is  the  velocity  of  sound  in  the  room,  and  wj,  ns,  and  n*  are  order  numbers  having 

values  of  Or  1,  2,  3, In  a  room  where  li  -  8  ft,  It  =  8  ft,  and  Z*  =  9.5  ft,  the  gravest 

mode  of  vibration  is  given  by  n±  =*  nj  =  0  and  ns  =  1 ;  that  is,  the  room  resonates  as  does 
an  organ  pipe,  9.5  ft  long  and  stopped  at  both  ends,  when  vibrating  at  its  gravest  mode. 
The  wavelength  of  this  gravest  normal  mode  is  19  ft,  and  tbe  frecpieaey  (assuming  c  = 
1125  ft  per  sec)  is  59.2  cycles.  Other  resonant  or  normal  frequencies,  in  ascending  order, 
for  this  room,  are  70.3,  92.8,  90.8,  116.0,  119.2  cycles,  and  continuing  in  a  triply  iofinite 
series"  of  frequencies  corresponding  to  increasing  values  of  the  »*s.  When  none  of  the  n's 
is  zero  the  resonant  standing  waves  are  oblique,  and  when  one  or  two  of  the  n's  are  aero 
the  waves  are  tangential;  that  isr  the  waves  move  parallel  and  gracing  to  one  or  two,  re- 
spectively, pairs  of  walls  in  the  rectangular  room. 

The  above-enumerated  normal  frequencies  for  the  room  under  discussion  have  been 
observed,  both  by  the  "reinforcement"  the  room  gives  to  steady  tones  of  these  frequencies, 
and  by  the  transient  decay  of  tones  which  have  approximately  these  frequencies.  Thus, 
Fig.  2  is  a  series  of  oscillograms  showing  the  free  decay  of  sound  for  this  room  when  it  is 
excited  with  tones  having  different  "driving"  frequencies  between  90  and  101  cycles,  as 
indicated  above  each  oscillogram.  When  the  room  is  excited  with  a  frequency  of  92.9 
cycles,  which  corresponds  closely  to  the  normal  mode  having  a  frequency  of  92.8  cycles, 
the  free  decay  is  made  up  almost  exclusively  of  this  one  mode,  and  its  decay  rate  is*  smooth 
and  almost  exactly  exponential.  Similarly,  when  the  room  is  excited  by  any  other  fre- 
quency which  differs  only  slightly  from  92.8  cycles,  the  free  decay  is  made  up  of  this  one 
mode,  and  the  frequency  during  decay  is  not  the  <$rm«g  frequency  but  is  always  ike  fre- 
quffncy  of  this  normal  mode.  When  the  room  is  excited  with  a  frequency  of  96.7  cycles, 
•vrhich  is  about  midway  between  the  frequencies  of  the  two  modes  of  92J3  and  99.&  cycles, 
the  free  decay  is  made  up  of  these  two  normal  modes,  which  are  about  equally  excited, 
giving  rise  to  the  pronounced  beats  of  about  7  per  second,  as  expected,  since  the  two  natural 
frequencies  differ  by  7  cycles.  When  this  same  room  was  excited  at  a  frequency  of  118 
cycles,  the  oscillogram  of  the  free  decay  revealed  the  coexistence  of  beats  at  about  3.3 
and  19.3  cycles,  which  no  doubt  arose  from  the  simultaneous  excitation  of  the  tliree  neigh- 
boring normal  frequencies  of  99.8,  116.0,  and  119.2  cycles.  No  matter  what  frequency  is 
used  for  exciting  the  room,  the  "reverberation"  always  consists  of  the  free  decay  of  ose 
Of  more  of  the  room's  normal  modes  of  vibration. 

The  foregoing  tesults  demonstrate  that  reverberation  is  not  the  persistence  ol  rays  of 
sound  which  continue*  after  the  source  is  stopped,  as  rays  of  sound  successively  rejected 
back  and  forth  in  the  room,  but  rather  is  the  free  decay  of  one  or  more  (and  usually  many) 
modes  of  vibration.  However,  Strutt  has  shown  that  the  forma!  law  governing  the  free, 
damped  vibration  of  the  sound  in  a  room  approaches  asymtotically  the  simple  reverbera- 
tion law  of  eq.  (2)  as  the  wavelength  of  the  exciting  sound  becomes  short  ia  comparison 
with  the  wavelength  of  the  gravest  mode  of  vibration  for  the  room.  Thus,  in  a  room 
(assuming  the  ab&orpifcive  material  to  be  fairly  well  distributed)  having  its  longest  dimen- 
sion 10  f t»  which  is  about  the  smallest  loom  in  widen  acoustics  is  a  factor  of  hnportaee, 


12-46 


ACOUSTICS 


the  wavelength  of  the  gravest  mode  is  20  ft.  In  such  a  room,  sound  having  a  wavelength 
of  one-tenth  of  that  of  the  fundamental,  namely,  2.0  ft,  would  be  sufficiently  short  to 
conform  reasonably  well  to  eq.  (2) .  ,  ^  T_  .L  .T. 

In  other  words,  when  the  room  is  filled  with  sound  having  a  wavelength  shorter  than 
2.0  ft,  that  is,  a  frequency  greater  than  about  560  cycles,  the  modes  of  vibration  which 
are  excited  are  so  numerous  and  their  frequencies  so  close  together  that  the  sound  in  the 
room  is  essentially  diffuse  and,  therefore,  the  requirements  are  fulfilled  for  the  approximate 
theories  of  reverberation  described  in  the  earlier  paragraphs  of  this  section,  especially  if 


FIG.  2.    Oscillograms  of  the  Decay  of  Sound  in  a  Small  Rectangular  Room,  Showing  that  the"Decay 
of  Sound  Consists  of  the  Damped  Free  or  Normal  Vibrations  of  the  Room 

the  absorptive  material  is  distributed  uniformly  over  the  boundaries  of  the  room  or  if 
some  means  are  provided  for  mixing  or  diffusing  the  sound  during  the  decay.  In  large 
rooms,  such  as  concert  halls,  church  or  school  auditoriums,  and  theaters,  the  lowest  modes 
of  vibration  are  usually  in  the  subaudible  range  of  frequencies,  so  that  the  elementary 
theory  of  reverberation  applies  with  adequate  rigor  in  such  rooms  for  all  frequencies  above 
about  100  cycles,  and  the  effects  of  room  resonance  usually  can  be  neglected.  (For  further 
details  consult  M.J.O.  Strutt,  Zeitschrift  angew.  Math.  Mech.;  Knudsen,  /.  Acous.  Soc. 
Am.,  Vol.  4,  20-37  [1932]  *  and  Morse  and  Bolt,  loc.  c#.) 

REVERBERATION  IN  COUPLED  SPACES.  When  two  or  more  inclosures  are 
coupled  by  means  of  openings  such  as  open  doors,  passageways,  or  even  thin  partitions 
which  are  capable  of  transmitting  an  appreciable  amount  of  sound,  the  reverberation  in 
each  inclosure  is  affected  by  the  reverberatory  properties  of  the  other  inclosure.  Thus, 
most  au-ditoriums  of  the  theater  type  are  divided  into  at  least  three  coupled  spaces — the 
stage,  the  main  portion  of  the  auditorium,  and  the  space  under  the  balcony.  Even  a 


REVERBERATION  AT  DIFFERENT  FREQUENCIES      12-47 

long  sound-recording  studio,  one  end  of  which  is  reverberant  and  the  other  end  non- 
reverberant,  may  be  regarded  as  two  coupled  spaces.  Office  space  is  often  divided  into  a 
number  of  coupled  spaces;  and  a  great  complexity  of  coupled  spaces  often  will  be  found 
in  cathedrals,  consisting  of  nave,  transepts,  choir,  sanctuary,  aisles,  chapels,  balconies, 
and  organ  chamber.  If  the  mean  free  path  can  be  determined  for  such  coupled  spaces,  and 
if  all  surfaces  have  approximately  the  same  absorption  coefficients,  the  regular  reverbera- 
tion formula,  using  the  appropriate  value  of  k,  will  give  the  time  of  reverberation  for  the 
entire  room.  (See  Knudsen,  Architectural  Acoustics,  Chapter  V.)  But  it  is  not  always 
feasible  to  determine  the  mean  free  path  for  a  complicated  combination  of  coupled  spaces, 
and  it  is  very  improbable  that  all  surfaces  will  have  even  approximately  the  same  coeffi- 
cients of  absorption.  In  many  instances,  therefore,  it  becomes  necessary  to  consider  the 
reverberation  in  each  of  the  several  coupled  spaces,  and  to  adjust  the  reverberation  time 
in  each  space  to  the  optimal  condition. 

Many  rooms  have  poor  acoustics  because  of  failure  to  recognize  the  effect  of  coupled? 
spaces.  Thus,  in  many  school  auditoriums  containing  a  balcony,  it  is  common  practice- 
to  install  nearly  all  the  absorptive  material  hi  the  ceiling.  In  some  auditoriums  the  walls 
under  the  balcony,  the  soffit  of  the  balcony,  and  the  floor  under  the  balcony  may  be  of" 
hard,  reflective  materials,  as  hard  plaster  and  concrete.  If,  in  addition,  the  seats  under 
the  balcony  are  of  the  unupholstered  type,  the  space  under  the  balcony  will  be  very  re- 
verberant although  the  space  in  the  main  part  of  the  auditorium  may  be  quite  free  from 
reverberation.  During  the  growth  or  decay  of  sound  in  such  an  auditorium  there  is  a 
transfer  of  energy  between  the  two  coupled  spaces,  with  different  rates  of  growth  or  decay 
in  the  two  spaces.  During  the  steady  state  the  rate  of  transfer  of  sound  from  the  dead 
space  to  the  live  one  is  equal  to  the  rate  of  transfer  in  the  opposite  direction,  During  the 
very  early  stages  of  the  decay  these  rates  of  transfer  are  nearly  equal,  but  since  the  sound 
decays'  much  more  rapidly  in  the  main  part  of  the  auditorium  than  it  does  under  the- 
balcony,  there  soon  will  be  established  an  excess  rate  of  flow  from  the  live  to  the  dead 
space,  and  the  result  is  that  the  reverberation  is  prolonged  in  the  main  part  of  the  audito- 
rium as  well  as  in  the  space  under  the  balcony. 

In  order  to  overcome  this  undesirable  condition  it  is  necessary  that  the  rates  of  decay 
in  both  spaces  be  nearly  equal  (or  that  the  rate  of  decay  in  the  smaller  space  under  the 
balcony  be  greater  than  the  rate  of  decay  in  the  main  part  of  the  auditorium).  This  in- 
volves a  determination  of  the  reverberation  in  both  spaces,  which  in  turn  necessitates  the 
assignment  of  coefficients  of  absorption  to  the  opening  which  couples  the  two  spaces.  It 
is  not  possible  to  assign  precise  coefficients  to  these  openings.  The  coefficients  will  depend, 
in  general,  upon  the  size  and  shape  of  the  opening,  the  depth  under  the  balcony,  and  the 
amoxmts  of  absorption  under  the  balcony  and  in  the  main  part  of  the  auditorium.  But  if 
botli  spaces  have  approximately  the  same  rates  of  growth',  as  they  should  for  good  acous- 
tics, the  "effective  coefficients"  will  be  of  the  order  of  0.40  to  O.SO — nearer  the  lower  limit 
for  shallow  recesses  which  contain  a  relatively  small  amount  of  absorption.  Similar  con- 
siderations apply  to  the  stage  opening  which  couples  the  stage  and  the  main  part  of  the 
auditorium. 

Many  theaters,  churches,  memorial  halls,  and  other  auditoriums  are  often  coupled,  by 
means  of  door  openings  or  archways,  to  rooms  or  corridors  which  are  excessively  rever- 
berant- In  such  auditoriums,  even  though  the  reverberation  in  the  audience  space  has 
been  adjusted  to  the  proper  value,  there  will  be  a  "feedback"  of  reverberation  from  the 
adjacent  reverberant  rooms  into  the  main  auditorium.  Thus,  auditors  in  a  theater  who 
a,re  seated  near  an  opening  to  a  reverberant  corridor,  foyer,  or  anteroom  will  be  disturbed 
by  the  excessive  reverberation  in  the  adjacent  room.  It  is  advisable  in  all  such  cases 
either  to  close  the  openings  or  to  use  an  adequate  amount  of  absorption  in  all  spaces  which 
are  coupled  to  the  audience  room.  In  general,  such  anterooms,  foyers,  or  corridors,  unless 
they  are  used  for  speech  or  music  rooms,  should  be  as  non-reverberant  as  possible. 

21.  REVERBERATION  AT  DIFFERENT  FREQUENCIES 

Unless  otherwise  specified,  it  is  generally  understood  that  the  time  of  reverberation  refers 
to  a  pure  tone  of  512  cycles.  Although  the  calculation  of  reverberation  at  a  single  fre- 
quency, as  512  cycles,  will  suffice  to  represent  the  reverberation  in  a  room  at  other  fre- 
quencies provided  the  absorptive  material  in  the  room  has  nearly  the  same  absorptivity 
at  all  frequencies,  or  provided  the  variation  in  absorptivity  is  known,  it  is  obvious  that 
such  is  not  the  case  if  the  absorptive  material  has  widely  different  and  undetermined  ab- 
sorptivities  at  different  frequencies.  Thus,  an  acoutical  plaster,  1/4  in.  thick,  applied  to- 
concrete  or  tile,  may  have  coefficients  of  absorption  of  0.06  at  128  cycles,  0.36  at  512  cycles,. 
/vnd  0.72  at  2048  cycles.  If  such  plaster  is  applied  to  the  entire  inner  surface  of  a  room 


12-48  ACOUSTICS 

the  reverberation  time  at  128  cycles  would  be  at  least  six  times  as  long  as  the  reverbera- 
tion  time  at  512  cycles,  and  the  time  at  2048  cycles  would  be  less  than  one-half  of  the  512- 
cycle  time.  If  the  reverberation  time  in  such  a  room  is  1.25  sec  at  $12  cycles,  it  will  be 
at  least  7.5  sec  at  128  cycles.  To  describe  this  room  as  one  having  a  time  of  reverberation 
of  1.25  sec — which  is  regarded  as  close  to  the  optimal  time  for  good  acoustics — certainly 
does  not  describe  the  reverberatory  properties  of  the  room ;  and  such  a  room  will  be  highly 
unsatisfactory.  There  will  be  complaints  of  excessive  reverberation,  and  the  room  will  be 
too  reverberant  for  the  bass  notes  of  music,  and  even  the  low-frequency  components  of 
speech  will  be  reverberant  and  overemphasized.  On  the  other  hand,  the  higher  tones  and 
harmonics  in  music  will  be  suppressed  owing  to  over-absorption  at  the  high  frequencies. 
Such  rooms  are  particularly  objectionable  for  recording  or  broadcasting  purposes. 

It  is  necessary  therefore  to  specify  and  calculate  the  reverberation  times  for  representa- 
tive frequencies  throughout  the  entire  range  used  in  speech  and  music.  It  will  be  found, 
however,  that,  if  calculations  are  made  at  128,  512,  and  2048  cycles,  the  resulting  times 
will  give  a  satisfactory  description  of  the  reverberatory  properties  of  the  room.  In 
recording  and  broadcasting  studios  it  is  desirable  to  consider  frequencies  as  high  as  8000 
cycles. 

In  general,  the  reverberation  time  at  128  cycles  should  be  slightly  longer  than  the  time 
at  512,  and  the  reverberation  time  above  512  should  remain  nearly  constant.  (See  Fig.  10, 
p.  12-75,  for  present  recommended  practice.)  The  success  or  failure  in  the  acoustical  de- 
sign of  rooms  will  depend  upon  the  selection  of  absorptive  materials  which  will  give  the 
pro-per  reverberatory  characteristic  throughout  the  entire  range  of  frequencies  used  in 
,-speech  and  music. 

22.  THE  MEASUREMENT  OF  REVERBERATION  AND  ABSORPTION 

COEFFICIENTS 

The  reverberation  time  of  a  room,  or  the  total  absorption  of  the  room,  for  rooms  in 
which  geometric  acoustics  will  apply,  can  be  determined  by  measuring  either  the  rate  of 
decay  of  sound  or  the  time  for  the  decay  between  known  intensity  limits.  For  determining- 
the  sound-absorptive  coefficients  of  materials  it  is  customary  to  make  measurements  ol  the 
rate  of  decay  of  the  sound  in  a  reverberant  room  first  when  the  room  contains  a  certain 
area  of  the  acoustical  material  to  be  tested  and  again  when  the  material  is  removed  from 
the  room.  In  order  to  approximate  conditions  which  wiEf  justify  the  applicability  of 
formulas  based  on  geometric  acoustics,  it  is  better  to  distribute  the  absorptive  material 
on  three  (non-parallel)  walls  rather  than  concentrate  in  one  area  on  one  walL  By  means 
of  eq.  (3),  the  value  of  5,  and  consequently  the  total  absorpjfckxn  aS,  can  be  calculated. 
The  absorption  of  the  acoustical  material  in  the  room  is  assuanaed  to  be  equal  to  the  dif- 
ference between  the  absorption  of  the  room  witli  the  material  m  it  and  the  absorption  of 
the  room  with  the  material  removed.  This  is  equivalent  to  assuming  that  <x  is  the  arith- 
metical mean  of  all  absorptive  surfaces  in  the  room,  an  assumption  which  is;  justifiable1 
provided  the  sound  in  the  room  is  kept  thoroughly  diffuse  duEriag  the  steady  state  and 
the  decay.  Warble  tones  at  least  100  cycles  in  breadth  with  a.  warble  frequency  of  at 
least  4  or  5  per  sec  should  be  used  for  test  tones  below  about  500  cycles,  and,  unless  the 
room  be  bounded  by  diffuse  reflectors,  large  rotating  vanes  shGHzkf  be  used  for  test  tones 
of  all  frequencies.  The  test  tones  should  be  pure. 

When  these  precautions  are  taken  the  rate  of  decay  will  confem  satisfactorily  to  the 
theoretical  rate,  and,  if  the  test  area  is  as  large  as.  about  72  sq  ft  im  a  room  having-  a  volume 
about  5000  to  10,000  cu  ft,  the  difference  between  the  rates  of  decay  with  and  without  the 
acoustical  material  in  the  room  will  be  large  enough  to  yield  coefficients  of  absorption 
accurate  to  about  =fcO.G5  for  frequencies  up  to  2000  cycles.  At  higher  frequencies  the 
absorption  in  the  air,  which  may  change  during:  the  time  required  for  the  completion  o£' 
the  test,  is  so  large  a  factor  that  errors  of  the  order  of  dbO.10  azee  isnavoidable  unless  the 
reverberation  room  is  carefully  air-conditioned.  Even  with  am  air-conditioned  room  the- 
accuraey  is  not  satisfactory  at  frequencies  above  4000  cycles,  because  the  absoarptioru  ib 
the  air  is  such  a  large  factor  that  the  difference  between  the  rates  of  decay  with  and  with- 
out the  acoustical  material  in  the  room  is  not  appreciable  nnTea&  the  test  area  JB  greatly 
increased. 

The  rate  of  decay  is  measured  by  some  type  of  reverberating  naeter  which  m  general 
coaasists.  of  (1)  a  suitable  source  of  steady  or  warble  tones,  usually  a  vacuum-tube  oscillator,, 
an  electrical  low-pass  filter,  a  power  amplifier,  and  an  eleetrodymamic  laudspeafce^;,  (2)  a, 
iMgh-quaEty  DMcrophone  and  amplifier;  (3)  an  electrical  attenuate  for  varying  the  e^fa* 
oC  ibe  ampfifier;  and  (4>  either  a  recorder  which  registers  comtBiuaasly,  on  a  moving,  pape^- 


MEASUREMENT  OF  REVERBERATION  AND  ABSORPTION      12-49 

chart  or  on  a  light-sensitive  medium,  a  graphic  record  of  the  decay,  or  some  type  of  indi- 
cator, usually  a  relay  and  chronograph,  by  means  of  which  the  rate  o£  decay  can  be  de- 
termined. 

The  Bell  Telephone  Laboratories,  Inc.,  among  others,  have  developed  a  high-speed  level 
recorder  which  gives  a  response  proportional  to  the  logarithm  of  the  actuating  current,  and 
the  instrument  is  so  adjusted  that  the  record  gives  directly,  when  the  paper  tape  is  moving 
with  constant  speed,  the  rate  of  decay  of  the  sound  in  decibels  per  second.  If  the  decay 
follows  the  exponential  law,  the  curves  will  be  straight  lines.  However,  since  the  decay 
consists  of  several  contiguous  frequencies  (normal  modes  of  vibration)  in  close  proximity 
to  the  frequency  of  the  exciting  tone,  there  wiH  be  interference  between  these  several 
frequencies  (each  of  which  decays  exponentially)  so  that  the  resultant  decay  curx-e  gen- 
erally will  be  quite  irregular.  Typical  decay  curves  obtained  with  this  instrument  in  the 
reverberation  room  of  the  Bell  Telephone  Laboratories  are  reproduced  in  Fig.  3.  As  these 


<Tinje  In  Seconds 


\ 

> 

/       \ 

^v     / 

\ 

/         ^ 

%I    / 

.   V 

/                 V 

4  i      —        i  " &  — T 

Time  in  Seconds 
FIG.  3.     Decay  Curves  Obtained  with  Bell  Laboratories  High-epeed  Level  Recorder 

records  show,  the  decay  is  not  strictly  exponential  but,  except  for  minor  fluctuations  which 
can  be  attributed  to  the  resonant  or  interfering  phenomena  discussed  in  article  20,  the 
general  trend  of  the  decay  conforms  very  satisfactorily  to  the  exponential  law,  over  a 
range  of  40  db;  and,  if  a  straight  line  is  fitted  to  the  recorded  curve  of  decay,  the  slope  of 
this  line  will  give  the  rate  of  decay  with  sufficient  accuracy  for  practical  purposes. 

The  curves  labeled  9,  10,  and  11  were  made  with  a  pure  tone  and  a  single  microphone 
and  with  the  recorder  adjusted  to  "follow"  maximal  speeds  of  decay  of  240,  120,  and  60 
db  per  sec,  respectively.  In  9,  for  example,  the  recorder  is  capable  of  following  the  actual 
decay  much  more  closely  than  it  is  in  11,  where  only  the  slower  variations  of  decay  are 
recorded.  The  curves  labeled  12,  13,  and  14  were  made  at  the  same  recorder  speeds,  re- 
spectively, but  a  warble  tone  was  used  instead  of  a  single  pure  tone.  The  advantage  of 
the  warble  tone  for  reverberation  measurements  is  obvious  from  these  decay  curves. 

In  the  chronographic  type  of  reverberation  meter,  in  use  in  many  laboratories.  It  is 
customary  to  measure  the  tune  for  a  given  drop  in  level,  increasing  the  drop  in  steps  of 
5  or  10  db.  A  good  meter  of  this  type  is  the  one  developed  by  F.  V.  Hunt  (/.  Acows.  Soc, 
Am.,  Vol.  5,  127  [1933],  and  Vol.  8,  34  [1936]),  which  is  almost  automatic.  As  usual,  the 
sound  source  is  a  warble  tone  oscillator.  Several  microphone  positions  throsighiOiit  t^e 
room  are  used  to  insure  a  good  average,  and  the  reverberant  sound  thus  detected  and 
amplified  is  rectified,  and  rapid  fluctuations  are  filtered  out.  An  automatic  timer  tunas 
off  the  source,  always  at  the  same  phase  of  the  sound  wave,  when  the  soimd  level  ia  tiae 
room  has  reached  a  predetermined  level,  indicating  the  time  for  this  decay.  This  is 
repeated  40  or  more  times,  and  the  average  value  is  used  for  plotting  the  <$ecay  earv^ 
Figure  4  is  a  composite  curve  obtained  by  Hunt,  showing  the  average  decay  for  a  warble 
tone  of  1000  ±  200  cycles  throughout  a  course  of  80  db.  By  comparing  the  observed 
decay  curve  with  the  dashed  straight  line,  it  will  be  seen  that  the  decay  is  satisfactorily 
exponential  during  the  first  50  db  of  decay,  the  portion  of  the  curve  which  should  be  tatsed 
in  making  sound-absorption  measurements.  The  non-linear  decay  from  50  to  80  dfe 
probably  results  from  the  more  slowly  damped  tangential  modes  of  vibration,  which  pre- 
dominate during  the  latter  stages  of  the  decay* 


12-50 


ACOUSTICS 


A  third  method  of  measuring  the  decay  rate  is  the  oscillographic,  illustrated  in  Fig.  2, 
which  gives  a  detailed  picture  of  the  frequencies,  as  well  as  the  intensity,  throughout  the 
course  of  the  reverberation.  . 

Reverberation  measurements  are  useful  not  only  for  determining  the  coefficients  of 
sound  absorption  of  acoustical  materials  in  a  reverberation  chamber  but  equally  for  deter- 
mining the  reverberatory  proper- 
ties of  all  rooms.  In  general,  a 
reverberation  meter  should  be 
capable  of  making  measurements 
at  all  frequencies  between  about 
128  and  4096  cycles  and  should 
reveal  the  detailed  nature  of  the 
decay  of  the  sound,  especially 
during  the  first  30  to  40  db  of  the 
decay.  In  music  rooms,  record- 
ing or  broadcasting  rooms,  and 
theaters,  it  is  desirable  to  make 
measurements  at  frequencies  as 
high  as  8000  cycles. 

For    additional    methods    for 


70 


60 


50 


DB 


30 


10 


N 

Bar 
Fou 
S^     Slope  o 

Composite 
e  Room  
r  Microphom 
F  Straight  Po 
from  Linear 

Decay  Curve 
1000  ±200 
>  Positions 
rtion—  23.6 
ty  0.006 

cps 
DB  Sec 

~^T*Ave  Dev 

N 

\ 

\ 

V 

FIG.  4. 


measuring  reverberation  and  ab- 
sorption, consult  Knudsen,  Archi- 
tectural Acoustics,  Chapter  VII; 
Sabine,  Acoustics  and  Architec- 
ture, Chapter  VI;  Olson  and 
Massa,  Applied  Acoustics,  Chap- 
Decay  Curve,  Based  on  Measurements  of  Average  ter  XII;  and  Beranek,  Acoustic 
Time  for  a  Given  Drop  in  Level.  (Hunt.)  Measurements. 


1 


2  3 

Time  -  Seconds 


23.  COEFFICIENTS  OF  SOUND  ABSORPTION 

In  the  following  charts  and  tables  there  is  given  a  fairly  complete  listing  of  the  coeffi- 
cients of  sound  absorption  of  the  materials  which  are  used  in  building  construction,  espe- 
cially for  acoustical  purposes.  Many  of  the  same  materials  have  been  measured  by  dif- 
ferent investigators,  and  the  results  are  not  always  in  good  agreement.  Such  factors  as 
actual  differences  in  the  samples,  differences  in  the  methods  of  measurement,  differences 
in  the  size,  shape,  and  location  of  the  samples  compared  with  the  size  and  shape  of  the 
test  rooms,  the  purity  of  the  test  tone  used,  and  errors  inherent  in  the  use  of  reverberation 
methods  and  formulas  based  on  geometric  acoustics  when  wave  and  not  geometric  acous- 
tics applies  are  probably  responsible  for  this  lack  of  agreement.  Where  large  discrepancies 
have  existed,  certain  liberties,  guided  by  experience  and  the  probable  influence  of  wave 
acoustics,  have  been  taken  in  averaging  results. 

Most  of  the  materials  manufactured  by  the  leading  acoustical  concerns  in  the  United 
States  have  been  tested  in  the  same  laboratory  by  the  same  method;  this  facilitates  com- 
parison of  different  materials.  The  authority  for  these  measurements  is  the  Acoustical 
Manufacturers'  Association;  the  tests  were  made  at  the  Riverbank  Laboratories.  The 
authorities  for  the  other  measurements  are  listed  in  Table  1. 

Much  progress  has  been  made  in  the  development  of  acoustical  materials  during  the 
past  decade,  and  this  progress  will  continue.  Many  improved  and  new  materials  will  be 
described  in  subsequent  issues  of  the  Bulletin  of  the  Acoustical  Materials  Association,  which 
is  published  at  frequent  intervals.  (These  bulletins  should  be  consulted;  they  can  be 
obtained  from  the  Acoustical  Materials  Association,  205  W.  Monroe  St.,  Chicago,  Illinois.) 

Table  1,  on  sound-absorption  coefficients,  includes  considerable  data  concerning  acousti- 
cal materials  in  addition  to  the  coefficients  of  sound  absorption,  such  as  the  type  of  the 
material,  the  nature  of  the  mounting,  the  light  reflection  coefficient  (usually  as  painted  by 
the  manufacturer),  the  weight  per  square  foot  of  the  material,  and  the  unit  size  of  the 
material.  The  numbers  used  to  describe  the  types  and  mountings  are  those  used  by  the 
Acoustical  Materials  Association.  The  legend  for  these  numbers  is  given  hi  the  table. 

In  choosing  materials  for  noise-reduction  purposes,  in  offices,  factories,  restaurants, 
hospitals,  etc.,  it  is  customary  to  rate  the  materials  acoustically  in  terms  of  the  arithmetic 
mean  of  the^  coefficient  at  256,  512,  1024,  and  2048  cycles.  This  average  value  is  called  the 
ywise-reduction  factor.  It  is  better  practice,  however,  to  choose  materials  that  have  the 
absorption  characteristics  best  adapted  to  the  characteristics  of  the  noise  to  be  absorbed 
It  is  especially  important  to  avoid  materials  having  very  low  absorption  at  the  low  fre- 
quencies for  sounds  which  are  made  up  predominently  of  low  frequencies. 


COEFFICIENTS  OF  SOUND  ABSORPTION 


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12-54 


ACOUSTICS 


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COEFFICIENTS  OF  SOUND  ABSORPTION 


12-55 


Hard  Plasters,  Masonry,  Wood,  and  Other  Standard  Building  Materials 

™m^mm™m 

—  —  *—  « 

-™—  _ 

**  Small  granules  of  exfoliated  verinioulite  bonded  with  a  planlio,  tacky  binder, 
ft  These  coefficients  aro  estimates  mado  by  the  compiler. 
Authority:  (1)  Bureau  of  Standards,  (2)  Acoufltioal  Materials  Aasuuwuou,  (3)  V.  O.  Knudaen,  (4)  P.  E.  Sabine,  (5)  W,  C.  Sabino,  (0)  Building  Research  Station,  (7) 
F.  R.  Watson,  (8)  Wente  and  Bedell,  (9)  average. 

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Audience,  mixed,  seated  in  theater  chairs,  single  padding  on  b 
Audience,  mixed,  seated  in  church  pews  
Chair,  American  logo,  full  upholstered  in  mohair  
Chair,  box  spring,  pantasote  seat  and  back,  plywood  on  rear;  a 
Chair,  plywood  seat,  plywood  back;  seats  up  ,  
Chair,  spring  edge  mohair  seat  and  back,  plywood  panel  on  rea 

ir  covered  seat  and  back 

"S       -       •    •   - 

Brick  wall,  unpainted.  . 

GlflAS  

Interior  stucco,  smooth  finish,  on  tile.  . 

MnrKlA  

Plaster,  gypsum,  on  hollow  tile  
Plaster,  gypsum,  scratch  and  brown  coa 
Plaster,  lime,  sand  finish,  on  metal  lath 
Poured  concrete,  unpainted  
Poured  concrete,  painted  and  varnishec 
Water,  as  in  swimming  pool  
Wood  sheathing,  pine  
Wood  veneer,  on  2  in.  by  3  in.  wood  st 

Chair,  like  above,  with  thick,  complete!: 
Chair,  theater,  heavily  upholatered.  .  .  . 

TWorm  «/JllH.  

seated  in  American  ioge 
high  school  
junior  high  school  
grammar  school  ,  . 

in,  without  coat,  seated.  . 
in,  with  coat,  aeated.  .  ,  .  . 

)    O    O    O    O    O    O    C 

12-56 


ACOUSTICS 


The  table  contains  absorption  data  on  a  number  of  mineral  wool  blankets  made  up  of 
wool  of  different  densities,  and  of  thicknesses  varying  from  1  to  3  in.  It  will  be  seen  that 
by  suitable  choice  of  density  and  thickness  it  is  possible  to  obtain  a  wide  variety  of  absorp- 
tion characteristics. 


1.00 

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Frequency— Cycles  per  Second 
PIG.  5.     Absorption  of  Different  Thicknesses  of  Hair  Felt.     (Wente  and  Bedell.) 

In  Fig.  5  are  shown  the  results  of  sound-absorption  measurements  of  Wente  and  Bedell 
on  hair  felt  of  different  thicknesses.  The  principal  effect  of  increasing  the  thickness  of  a 
porous  material  is  to  increase  the  absorption  at  the  low  frequencies. 

In  Pig.  6  are  shown  the  results  of  measurements  at  512  cycles  on  different  thicknesses 
of  hair  felt  obtained  in  four  different  laboratories.  These  results  show  not  only  the  effect 
of  thickness  but  also  the  order  of  agreement  of  measurements  made  at  different  laboratories. 

The  manner  of  mounting  acoustical  materials  influences  their  absorptivity.  Most 
fibrous  and  porous  materials  increase  in  absorptivity,  especially  at  frequencies  below  500 
cycles,  as  the  thickness  of  the  air  space  behind  the  material  increases.  Thus,  fiber  board 


l.UU 

g.SO 

CM 

S.60 

c 

V 

i 

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i 

5= 

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/ 

f 

Wf 

I.  Sabine 
.Wente 
Paris 
.  Knudsen 

P    T 

/ 

/ 

)                     1.0                    2.0                   3.0                  4.0                   5.0                   6*< 

Thickness  of  Hairfelt-foches 
Fi<3.  6,    Absorption  of  Hair  Felt  as  Determined  in  Different  Laboratories 

and  tiles,  acoustical  plaster,  and  similar  materials,  are  more  absorptive  at  low  frequencies 
when  furred  out  from  a  dense,  rigid  wall  than  when  applied  directly  to  the  wall. 

The  absorp*ivities  of  such  materials  as  acoustical  plaster  are  dependent  upon  the 
composition-  as  well  as  upon  the  manner  of  applying  and  drying.  If  too  much  binder 
material  is  used  the  plaster  is  not  sufficiently  porous;  if  an  insufficient  amount  of  binder 


NOISE  MEASUREMENTS  12-67 

is  used  the  plaster  does  not  set  hard.  Likewise,  if  the  undercoats  of  plaster  are  too  wet 
(or  "green"),  the  binder  material  forms  an  impenetrable  film  at  the  surface,  whereas  if 
the  undercoats  are  too  dry  the  binder  material  is  absorbed  by  the  undercoats,  and  the 
plaster  will  crumble.  In  order  to  obtain  good  results  with  acoustical  plaster  it  must  be 
applied  by  competent  plasterers  in  strict  conformity  with  the  specifications  of  responsible 
manufacturers . 

The  absorptivity  of  acoustical  plaster  or  fiber  board  may  be  ruined  by  decoration  with 
oil  or  water  paint,  varnish,  distemper,  or  other  materials  which  will  close  the  surface  pores. 
Viscous  or  heavy  paints  which  bridge  over  or  dose  the  surface  pores  must  be  avoided. 
Such  materials  as  thin  aniline  dyes,  gasoline  or  kerosene  stains,  thin  lacquer  sprays,  or 
dry  paint  dusted  on  with  a  pounce  bag  are  satisfactory  means  of  decoration  without  im- 
pairing the  absorptive  properties  of  plaster  or  fiber  board.  Certain  commercial  materials 
containing  large  holes,  or  materials  covered  with  a  perforated  facing,  may  be  decorated  with 
lead  or  oil  or  any  other  kind  of  paint,  provided  the  paint  does  not  bridge  over  the  holes. 

In  order  to  facilitate  convenient  use  of  the  table  the  materials  have  been  grouped  as 
follows:  acoustical  tiles,  boards,  and  sheets;  acoustical  plasters  and  sprayed-on  plastic 
materials^  mineral  wool,  Fiberglas  blankets,  and  acoustical  felts;  set  materials  for  use  in 
motion-picture  studios;  hangings,  floor  coverings,  and  miscellaneous  materials;  hard 
plasters,  masonry,  wood,  and  other  standard  building  materials;  and  audience,  individual 
persons,  chairs,  and  other  objects.  The  tabulation  in  each  group  of  materials  is  arranged 
in  alphabetical  order. 

24.  PRACTICAL  CONSIDERATIONS  OF  SOUND-ABSORPTIVE 

MATERIALS 

In  making  a  choice  of  absorptive  materials  a  number  of  other  factors  must  be  considered 
besides  the  coefficients  of  sound  absorption.  Good  acoustics  is  only  one  of  many  qualities 
which  should  be  secured  in  every  building.  Thus,  besides  absorption  coefficients,  it  is 
necessary  to  consider  such  factors  as  the  following:  structural  strength;  decorative  pos- 
sibilities; adaptability  to  the  surface  available  for,  or  requiring,  absorptive  treatment; 
maintenance;  sanitation;  ease  of  application;  fire  hazard;  absorption  of  water;  attraction 
for  vermin;  "fool-proof ness";  durability;  and  cost.  Each  room  requires  a  certain  amount 
of  sound  absorption;  certain  surfaces  in  some  rooms  require  highly  absorptive  treatment. 
These  two  conditions  usually  limit  the  choice  of  absorbents  to  materials  having  coeffi- 
cients within  certain  specified  limits.  As  a  rule,  however,  many  materials  having  coeffi- 
cients within  these  limits  will  be  available.  This  allows  considerable  freedom  hi  the  selec- 
tion of  materials  which  will  be  satisfactory  not  only  acoustically  but  also  for  the  other 
requirements.  For  a  detailed  discussion  of  the  choice  of  sound-absorptive  materials 
consult  any  standard  textbook  on  architectural  acoustics  (as  Knudsen  and  Harris,  Acous- 
tical Designing  in  Architecture). 

For  bibliography,  see  p.  12-76. 

SOUND  INSULATION 

By  Vern  O.  Kncdsen 

25.  NOISE  MEASUREMENTS 

During  recent  years,  acoustical  engineers  and  civic  authorities  have  become  increasingly 
aware  of  the  problems  associated  with  the  measureme^it  and  abatement  of  noise.  One  of 
the  prime  requirements  for  good  acoustics  in  every  room  is  absence  of  noise,  i.e.,  unwanted 
sound.  In  the  design  of  theaters,  music  rooms,  churches,  schools,  office  and  industrial 
buildings,  hotels,  apartment  houses,  and  studios  for  the  recording  or  broadcasting  of 
sound,  it  is  necessary  that  the  designing  architect  or  engineer  know  (1)  the  amount  aad 
kind  of  the  noise  against  which  he  is  to  provide  insulation,  and  (2)  the  amount  of  noise 
which  can  be  tolerated  in  different  types  of  buildings.  The  difference  between  the  mag- 
nitudes of  (1)  and  (2)  gives  the  amount  of  sound  insulation  which  should  be  provided  in 
the  building.  As  -discussed  in  article  5  of  this  section  the  magnitudes  of  noises  vary  over 
a  wide  range. 

The  magnitude  and  character  of  steady  sounds  can  be  represented  by  plotting  the 
intensity  level  (per  cycle  or  for  a  specified  band  width)  as  a  function  of  the  sound  frequency. 
In  Fig.  1,  the  intensity  level  per  octave,  i.e.,  the  intensity  levels  as  measured  in  octave 
bands,  for  a  number  of  ordinary  sounds  are  plotted  for  the  important  audio-frequency 
range  of  50  to  10,000  cycles.  Traffic  noise  will  be  seen  to  have  its  predominant  intensity 


12-58 


ACOUSTICS 


at  low  frequencies;  the  noise  from  typewriters,  on  the  other  hand,  is  greatest  at  high  fre- 
quencies. Such  characteristics  of  noise  should  be  considered  in  the  problems  of  sound 
insulation  and  noise  reduction  in  building  design. 

Most  measurements  of  noise,  of  interest  in  building  design,  have  been  made  with  com- 
mercial sound-level  meters  which  measure  the  overall  sound  level  in  decibels  rather  than 
the  intensity  level  as  a  function  of  the  frequency.  The  latter  should  be  used  whenever 
available,  but  the  sound  level  corresponds  roughly  with  the  sensation  of  the  sound  and 
provides  a  convenient  numerical  scale  for  comparing  the  levels  of  different  sounds.  Thus, 
Fig.  2  gives  the  average  sound  levels,  in  decibels,  as  measured  with  a  sound-level  meter, 
for  ajlarge  number  of  locations  in  or  near  buildings. 
90, 


8  " 

1570 


1^60 


i  50 


40 


^ 


1  1  I  1 


^ 


\     \ 


V 


T    1    I  1 


50 


1OO 


200 


5,000        10,000 


500  1,000         2,000 

Frequency  (cycles  per  second) 

FIG.  1.     Sound  Spectra  of  Some  Typical  Noises.     (Fleming  and  Allen.)     (British  Crown  Copyright 
reserved.    Reproduced  by  permission  of  the  Controller  of  His  Britannic  Majesty's  Stationery  Office.) 

Examples  of  Noise  Analysis 

A — Traffic  noise.    Average  of  miscellaneous  vehicles  passing  at  about  20  ft. 
B — Typists'  room.    Two  typewriters  in  operation. 
C — Woodwork  shop.    14-in.  circular  saw. 
D — Woodwork  shop.    Planing  machine. 

If  a  sound  wave  completely  modulates  the  pressure  of  the  air  at  sea  level,  i.e.,  if  the 
instantaneous  pressure  varies  from  0  to  2  atmospheres,  the  intensity  level  would  be  194  db. 
This  represents  an  upper  possible  limit  for  the  intensity  of  sounds,  a  limit  which  is  not  far 
above  that  actually  attained  with  the  Victory  Siren  (a  large  air  stream  modulated  with  a 

1  'chopper"),  which  was  used  in  New  York  City  and  other  American  cities  as  an  air-raid 
alarm  during  World  War  II.    Most  of  the  other  entries  in  Fig.  2  are  self-explanatory.    The 
standard  deviations  are  given  for  a  number  of  measurements;  these  were  made  by  Bell 
Telephone  Laboratories  (D.  F.  Seacord,  J.  Acoits.  Soc.  Am.,  Vol.  12,  183)  at  more  than 
600  locations  in  four  different  cities.     The  measurements  hi  the  private  hospital  room 
(made  by  the  author  with  a  continuous  recorder)  revealed  sound  levels  of  50  to  58  db  dur- 
ing 80  per  cent  of  the  time  from  5:00  P.M.  to  7:00  P.M.;  at  9:00  P.M.  the  level  had  dropped 

2  or  3  db.    In  the  corridors  of  this  same  hospital,  levels  of  65  to  75  db  were  common;  there 
were  peak  levels  of  78  db  from  the  closing  of  elevator  doors  55  ft  away  from  the  sound-level 
recorder,  and  90  db  from  coughing  40  ft  away. 

26.  ACCEPTABLE  NOISE  LEVELS  IN  DIFFERENT  BUILDINGS 

Although  it  is  not  customary  for  building  codes  to  specify  the  allowable  noise  in  dif- 
ferent types  of  buildings,  and  opinions  differ  considerably  as  to  tolerable  noise  levels,  the 
following  table  gives  approximate  loudness  levels  which  will  be,  in  general,  highly  satis- 
factory: 

Decibels 

Radio,  recording  and  television  studios 25  to  30 

Hospitals "  "    35  to  40 

Music  rooms 30  to  35 

Apartments,  hotels,  and  homes 35  to  45 

Theaters,  churches,  auditoriums,  classrooms,  and  libraries 30  to  40 

Private  offices  and  conference  rooms 35  to  45 

Large  public  offices,  banking  rooms,  stores,  etc 45  to  55 

Restaurants 50  to  55 

Factories 45  to  80 


NOISE  LEVELS  IN  DIFFERENT  BUILDINGS  12-59 


Special  conditions  or  circumstances,  such  as  past  experience,  other  near-by  noises,  and 
costs,  may  alter  the  acceptable  noise  levels,  but  the  levels  given  in  the  table  are  recom- 
mended for  purposes  of  building  design.  As  will  be  seen  by  comparing  these  values  with 
those  given  in  Fig.  2  the  acceptable  values  given  in  the  table  are  somewhat  lower  than 


Noise  Out-of-doors 

Deci- 
bels 

Noise  in  Buildings 

30  ft  from  Victory  Siren  (assuming  a  50-kw 
sound  source  and  n<>  ft+tftrnijtflon)                   > 

-200- 

Complete  modulation  of  air  pressure  at  sea  level 

Test  chamber  for  airplane  motors  (l50Q-hp* 
propeller  type) 

Very  noisy  electric  power  substation 

Subway  station,  express  train  passing 
Electric  power  sjubstatkm  (about  average) 

Ventilating  and  eqalpment  room  for  large  note! 

Average  factory  (standard  deviation  =12  db) 
Kaiser  William  Memorial  Church..  Berlin  (usuaj 
daytime  traffic) 
Large  offices  (standard  devIailon—4.5  db) 
Large  stores  (standard  deviations*:  6.0  db) 

Private  room,  hospital,  on  WUshlre  Boulevard, 
Los  Angeles 
Average  residence,  with  radio  (standard  deviation— 
,  8.0  db) 
Average  residence,  without  radio  (standard  delation 
-5.5  db) 
Average  church 

Theater,  no  audience,  quiet  location 
Quiet  suburban  residence,  at  night,  no  near-by 
traffic 

Quiet  sound  studio  for  making  motion  pictures 

Reverberation  room,  University  of  California 
at  Los  Angeles 

-180— 

—160- 

—120- 

Nolse  from  4-motor  transport  alcplane,  •  > 
2000  ft  overhead 

Heavy  automobile  traffic  on  WHshke  Boulevard,  —  >• 
Los  .Arjgeles 

—lOCh^ 

—80  — 

—60— 

HoJIywood  Bowl,early  evening,  no  audience  present-*- 

=  4O— 

—20— 

—  0  — 

FIG.  2.     Sound  Levels  in  or  Near  Buildings 

those  that  prevail  in  existing  buildings.  The  average  loudness  level  in  the  private  hospital 
room,  for  example,  is  54  db,  which  is  some  15  to  20  db  greater  than  the  level  proposed  in. 
the  table.  Since  the  outside  traffic  noise  adjacent  to  this  hospital  has  an  average  level, 
when  traffic  is  heavy,  of  80  db,  and  since  the  average  tolerable  level  for  a  hospital  is  37  db, 
the  building  should  be  designed  for  sound  insulation  in  such  a  manner  as  will  provide  an 
overall  noise  reduction  80-37  db,  i.e.,  43  db.  Part  of  the  required  noise  reduction  can  be 
accomplished  by  a  proper  "setback"  of  the  building  and,  in  some  instances,  by  dense 
planting  of  evergreen  shrubs  and  trees  between  the  street  and  the  building;  another  part, 
by  the  use  of  sound-absorptive  materials  within  the  building;  but  most  of  the  reduction 
can  be  accomplished  only  by  proper  sound  insulation  of  the  building  itself. 


12-60 


ACOUSTICS 


27.  FUNDAMENTAL  PRINCIPLES  OF  SOUND  INSULATION 

Nearly  every  building  is  subject  to  the  annoyance  of  noises  which  have  their  origin  in 
adjacent  rooms  or  outside.  It  is  possible  to  design  buildings  in  such  a  manner  as  to  ex- 
clude effectively  both  these  types  of  annoying  noises.  The  principal  means  whereby  such 
noises  enter  a  building  are  the  following:  | 

1.  By  means  of  openings,  as  windows,  cracks  around  doors,  ventilating  ducts,  or  any 
other  openings  that  will  admit  a  free  flow  of  air. 

2.  By  means  of  refraction  or  transmission  through  partitions.    This  is  analogous  to  the 
refraction  or  transmission  of  light  from  air  to  water,  or  between  any  other  two  dissimilar 
media. 

3.  By  means  of  the  conduction  of  sound  through  solids.    For  example,  "impact  sounds," 
such  as  footfalls,  hammering  on  walls  or  floors,  or  the  moving  of  furniture  on  hardwood 
floors,  are  conducted  through  the  dense  and  rigid  structural  members  of  a  building. 

4.  By  means  of  the  diaphragm  action  of  walls  which  communicate  sound  from  one  side 
of  a  partition  to  the  other  side. 


.l/'-J-M 
Isolation  Felt 


One  Design  of 

Johns-Manvllle 

Floor  Chair 


Furring  Channel 


.Metal  Latb 
Plaster 


J-M  Wall 
Isolator 


.Tie  Wires 
•Wood  Ground 
-Wood  Sleeper 
I-M  Floor  Chair 


Typical  JohnszManville  Wall  Isolating  Treatment 


Typical 

Johns-Manville 
Ceiling  Isolator 


As  Used  by  United  States  Gypsum  Co. 
FIG.  3.    Flexible  Cushions,  Supports  and  Connectors 

The  refraction  or  transmission  of  sound  from  one  medium  to  another,  as  from  air  to 
plaster  or  stone,  is  an  almost  negligible  factor  in  building  construction — not  more  than 
about  one-millionth  of  the  intensity  of  the  incident  wave  in  air  can  penetrate  a  material 
like  brick  or  stone. 

The  transmission  of  sound  through  openings,  on  the  other  hand,  is  often  the  means  by 
which  sound  is  most  readily  transmitted  from  one  portion  of  a  building  to  another.  This 
means  of  transmission  often  limits  the  amount  of  sound  insulation  which  can  be  obtained 
in  buildings,  especially  in  hotels  and  apartments  where  the  insulation  is  determined  by 
such  unavoidable  openings  as  may  be  incidental  to  the  use  of  windows  and  doors.  Under 
such  circumstances  it  would  be  futile  to  provide  a  relatively  high  insulation  through  the 
se|>&rating  walls  or  partitions.  Even  very  small  openings,  such  as  cracks  around  doors 
or  around  imperfectly  fitting  windows,  are  effective  in  transmitting  a  considerable  amount 
of  sound. 


FUNDAMENTAL  PRINCIPLES  OF  SOUND  INSULATION      12-61 


The  transmission  of  sound  through  ventilating  ducts  often  becomes  &  troublesome  prob- 
lem. There  are  three  types  of  sound-transmission  which  must  be  controlled:  (1)  the  noise 
from  the  fans,  motors,  and  other  air-conditioning  equipment  which  is  transmitted  through 
the  ducts  and  into  the  room;  (2)  the  noise  from  an  adjacent  room  which  is  transmitted 
from  opening  to  opening,  ^  of  ten  through  a  short  and  highly  conductive  section  of  duct; 
and  (3)  the  noise  from  adjacent  rooms-  or  outside  which  may  be  transmitted  through  the 
walls  of  the  duct  and  thence 
through  the  ducts  and  into  the 
room.  The  noise  from  the  ven- 
tilating equipment  room  can  be 
reduced  suitably  by  (1)  the 
selection  of  slow-speed,  quietly 
operating  equipment;  (2)  treat- 
ing the  walls  and  ceiling  of  the 
equipment  room  with  highly 
absorptive  material;  and  (3) 
introducing  acoustical  attenua- 
tion within  the  ducts,  which 


W/y/w#2&3^^ 

M 


can  be  accomplished  by  using  FlG*  4*    m***™  Method  for  Insulation  of  Vibration 

very  long  ducts  of  small  cross-sectional  area  and  by  lining  the  ducts  with  highly  absorptive 
material,  or  by  introducing  other  acoustical  filters  in  the  ducts. 

Solid-borne  sound  travels  through  the  structural  members  of  a  building  with  but  very 
little  attenuation.  The  compressions!  wave  in  a  solid  is  often  communicated  to  large 
surfaces,  as  the  walls  or  floor  of  a  room,  and  these  large  surfaces  are  made  to  vibrate  like 
the  sounding  board  of  a  piano.  In  this  way  a  large  portion  of  the  solid-borne  sound  may 
be  radiated  into  a  room.  Many  solid-borne  sounds  can  be  controlled  by  the  carpeting  of 
floors;  by  the  wrapping  of  pipes — especially  where  they  touch  the  frame  of  the  building — 
with  flexible,  porous  materials,  as  hair  felt;  or  by  the  proper  mounting  of  machinery  on 
flexible  or  elastic  supports,  so  that  the  natural  frequency  of  the  machinery  mounted  on  its 
flexible  support  will  be  low  in  comparison  with  the  frequencies  which  are  to  be  insulated. 
One  of  the  most  effective  methods  of  eliniinating  these  solid-borne  sounds  is  to  introduce 
discontinuities  in  the  paths  of  the  conducted  sounds.  These  discontinuities  should  consist 
of  materials  which  differ  largely  in  elasticity  and  density  from  the  solid  structure  of  the 
building.  For  example,  it  is  possible  to  suspend  the  ceiling  by  means  of  flexible  supports; 
it  is  possible  to  build  up  inner  walls  in  a  room  which  are  fastened  to  the  monolithic  frame 
by  means  of  flexible  ties;  and  it  is  possible  to  float  the  floor  of  the  room  upon  flexible  pads 
of  cork  or  felt  or  other  elastic  material.  In  Fig.  3  are  shown  two  types  of  flexible  cushions, 
supports,  and  connectors  which  are  effective  for  insulating  solid-borne  vibrations  in 
buildings.  Figure  3  also  shows  some  constructional  details  for  insulating  solid-borne  sound 
in  buildings. 

Figure  4  shows  a  simple  method  of  insulating  any  object,  as  a  part  of  a  building  or  a 
piece  of  equipment,  from  earth,  building,  or  machinery  vibration.  The  problem  consists 
of  insulating  a  mass  TO  from  another  object  of  mass  M  by  means  of  a  flexible  support 
which  has  certain  elastic  and  damping  properties,  Figure  5  is  the  electric  circuit  equivalent 

M 


o 
o 
o 

o  ^^ 

o 

o 
o 


FIG.  5.     Electric  Circuit  Equivalent  of  Fig.  4 

of  Fig.  4.  This  circuit  implies  that,  when  M  is  set  into  forced  periodic  vibration,  these 
vibrations  are  communicated  to  m  principally  by  means  of  the  elastic  coupling  between 
M  and  m,  although  the  internal  damping  or  resistance  r  of  the  system  also  contributes  to 
the  coupling.  If  ai  and  03  represent  the  amplitudes  of  vibration  set  up  in  M  and  m, 
respectively,  n  the  frequency,  and  c  the  compliance,  then 


*  -f 


[2irmn  —  (l/2irnc)P 


(1) 


12-62 


ACOUSTICS 


This  equation  has  been  tested  experimentally  for  both  supported  and  suspended  systems 
and  is  in  good  agreement  with  the  observed  results.  The  equation  is  useful  for  calculating 
the  insulation  value  of  different  types  of  flexible  supports.  For  values  of  n  which  are 
small  compared  with  the  natural  or  free  vibration  of  m  upon  its  elastic  support,  a2M  will 
be  equal  to  unity;  that  is,  m  and  M  vibrate  with  the  same  amplitude.  At  the  resonant 
frequency,  a2/ai  is  greater  than  unity,  or  the  insulating  support  actually  amplifies  the 

motion  imparted  to  m.     However,  at  frequencies  greater  than  —  Vmc  the  value  of 


becomes  less  than  unity,  and  it  approaches  the  value  r/Z-n-nm  at  frequencies  which  are 
high  compared  with  the  natural  or  resonant  frequency.  In  general,  both  m  and  c  should 
be  as  large  as  possible  if  m  is  to  be  well  insulated  from  the  vibrations  in  M;  that  is,  the 
support  should  be  very  elastic  and  loaded  as  heavily  as  possible  (see  also  article  30)  . 

In  Table  1  are  given  the  values  of  the  compliance  c  and  the  resistance  r  of  a  number  of 
materials  which  are  used  for  the  insulation  of  vibration.  Beside  these  materials  there  are 
a  number  of  patented  devices,  similar  to  those  shown  in  Fig.  3,  which  are  very  effective. 
(See  Sound  Transmission  in  Buildings,  by  Fitzmaurice  and  Allen,  His  Majesty's  Stationery- 
Office,  London  [1939],  and  Modern  Theory  and  Practice  in  Building  Design,  by  Fleming 
and  Allen,  The  Institution,  London  [1945].) 

Table  1.    Compliance  and  Resistance  Data  for  Typical  Specimens  of  Flexible  Materials 

The  compliance  and  resistance  given  in  the  table  are  for  specimens  1  in.  thick  and  1  sq  cm  in  cross- 

section. 


Material 

Description  of 
Material 

Approximate 
Upper  Safe 
Loading,  Ib 
per  sq  in. 

Compliance, 
c,  cm  per 
dyne 

Resistance, 
r,  absolute 
units 

Corkboard                            

1.  10  Ib  per  board  ft 

12 

0.25  X  10~6 

0.15  X  105 

0.70  Ib  per  board  ft 

8 

0.50  X  10~6 

0.25  X  105 

1  .  35  Ib  per  board  ft 

4-6 

0.60  X  10~6 

0.50  X  10s 

Celotex                 

Insulating  board 

12 

0.18  X  10~6 

Insulating  Hr»ar(J 

15 

0.16  X  10~6 

^lasonite       .  .                         .  «  . 

Tnsnlfiting  Tx>ftT-d 

15 

0  1  2  X  1  0~6 

Sponge  rubber          .  .          

25  Ib  per  cu  ft 

1-3 

3.0  X     10~6 

55  Ib  per  cu  ft 

3-6 

1.2    X  10~6 

Hair  felt  

1  0  Ib  per  cu  ft 

'1-2 

1.5    X  10"6 

In  the  choice  of  materials  for  the  insulation  of  vibration  or  solid-borne  sound,  it  is 
necessary  to  give  consideration  to  the  safe  amount  of  loading  the  material  will  withstand 
without  breaking  down  or  without  being  compressed  to  the  extent  that  its  compliance  is 
reduced  beyond  required  limits.  It  also  is  important  to  select  a  material  that  will  have  a 
long  life  and  that  will  not  continue  to  compress  or  settle  under  the  load  which  it  supports. 
For  example,  if  ordinary  insulation  cork  is  loaded  as  much  as  20  or  30  Ib  per  sq  in.,  the 
material  will  continue  to  compress  indefinitely,  and  at  the  same  time  will  become  less 
and  less  compliant,  until  ultimately  it  not  only  loses  its  insulation  value  but  also  allows 
an  amount  of  settling  which  cannot  be  tolerated.  For  example,  a  specimen  of  1-in.  insula- 
tion cork  (0.70  Ib  per  board  ft),  under  a  load  of  20  Ib  per  sq  in.,  settled  0.04  in.  during  the 
first  24  hours,  0.02  in.  during  the  next  24  hours,  and  0.11  in.  during  the  first  5  months  it 
was  under  compression.  The  same  specimens,  under  a  load  of  10  Ib  per  S&  in.,  settled  only 
0.01  in.  during  the  first  24  hours,  0.005  in.  during  the  next  24  hours,  and  only  0.03  in. 
during  the  first  5  months.  In  general,  the  most  satisfactory  material  will  be  one  that  has 
a  high  compliance  and  very  little  tendency  to  settle  under  the  influence  of  the  load  and 
that  tends  to  return  to  its  initial  condition  when  the  load  is  removed.  Hair  felt,  cork,  and 
rubber  seem  to  be  the  best  available  materials  that  meet  these  requirements,  although  all 
these  materials  continue  to  settle,  and  become  less  and  less  compliant,  as  they  become 
older.  Flexible  steel  supports  and  clips,  such  as  those  shown  in  Fig.  3,  do  not  have  these 
defects  and  are  proving  to  be  very  satisfactory  not  only  for  the  insulation  of  walls,  floors, 
and  ceilings  but  also  for  insulating  all  sorts  of  equipment  from  the  floor  or  the  rigid  frame 
of  the  building. 

INSULATION  OF  SOUND  BY  POROUS  MATERIALS.  The  insulation  of  sound  by 
porous  materials  is  accomplished  principally  by  viscous  losses  within  the  capillary  pores 
within  the  material  and  by  the  vibration  of  the  component  parts  of  the  material.  Fig- 
ure 6  shows  the  transmission  coefficients  at  different  frequencies  for  one,  two,  three,  and 
four  layers  of  hair  felt  having  a  density  of  12  Ib  per  cu  ft.  These  results  show  that,  approxi- 
mately, the  logarithm  of  the  energy  reduction,  or  the  reduction  in  decibels,  of  sound 


FUNDAMENTAL  PRINCIPLES  OF  SOUND  INSULATION      12-63 


0-5 


02 


o-i 


OO5 


transmitted  through  porous  materials  is  proportional  to  the  thickness  of  the  material. 
The  results  also  show  that  porous  materials,  if  used  by  themselves,  do  not  provide  a  very 
high  degree  of  sound  insulation  unless  the  insulating  blanket  or  partition  is  very  thick. 
Thus,  at  700^ cycles,  the  coefficient  of  transmission  for  four  layers  of  hair  felt— each  layer 
is  0.58  in.  thick — is  about  0.01;  that  is,  a  sound  wave  of  700  cycles  would  be  attenuated 
only  20  db  in  passing  through  2.32  in.  of  hair  felt.  However,  when  such  materials  are 
used  properly  in  conjunction  with  rigid  partitions,  they  may  contribute  a  considerable 
amount  to  the  total  insulation  sup- 
plied by  a  wall  structure.  One  of  I -Or 
the  most  effective  ways  in  which 
such  materials  may  be  used  for  the 
insulation  of  sound  is  by  suspending 
or  supporting  the  porous  blanket  in 
an  air  space  between  two  rigid  par- 
titions. 

INSULATION  OF  SOUND  BY 
RIGID  PARTITIONS.  Sound  is 
transmitted  through  rigid  partitions 
principally  by  the  forced  vibration 
of  the  wall;  that  is,  the  entire  parti- 
tion is  forced  into  vibration  by  the 
pressure  variations  of  the  incident 
sound  wave.  The  transmission  co- 
efficient T  for  a  heavy  partition,  as 
masonry  or  concrete,  for  normally 
incident  sound,  is  given,  approxi- 
mately, by 


0-02 


o-oi 


where  pi  and  ci  are  the  density  and 
sound  velocity  in  the  partition,  p 
and  c  the  corresponding  density  and 
velocity  in  the  air,  ki  =  STT/XI,  where 
Xi  is  the  wavelength  of  the  sound  in 
the  partition  and  I  is  the  thickness 
of  the  panel.  The  transmission  loss 
in  decibels  (10  logio  1/r)  for  a  9-in. 
brick  wall,  as  calculated  by  eq.  (2), 
is  about  50  db  at  100  cycles  and 


0-005 


0-ooz 


O-OOl 


400 


1200 


1600 


600 
Frequency 

FIG.  6.     Transmission  Coefficient  for  One,  Two,  Three, 
and  Four  Layers  of  TTflir  Felt.    (Davis  and  Littler.) 


rises  almost  uniformly  with  frequency  to  78  db  at  4000  cycles,  and  then  drops  to  a 
minimum  at  8000  cycles  (which  is  zero  according  to  eq.  [2]  but  actually  is  of  the  order  of 
50  db  because  of  dissipation  within  the  partition).  At  frequencies  below  about  100  cycles, 
the  resonant  properties  of  the  partition  are  of  considerable  influence;  the  transmission 
loss  (T.L.)  at  these  low  frequencies  may  be  much  less  than  that  calculated  by  eq.  (2). 
The  observed  T.L.  for  heavy  masonry  partitions,  even  at  100  to  4000  cycles,  is  some  10  to 
15  db  less  than  that  calculated  from  eq.  (2),  but  the  equation  is  useful  for  predicting  the 
influence  on  T.L.  of  such  factors  as  the  wavelength  of  the  sound  and  the  mass  and  thick- 
ness of  the  partition. 

For  most  rigid  walls  in  buildings,  as  concrete,  brick,  clay  or  gypsum  tile,  wood  or  metal 
studs  plastered  on  one  or  both  sides,  and  even  glass  or  metal  panels,  eq.  (2)  can  be  further 
simplified  so  that,  approximately,  for  frequencies  of  100  to  4000  cycles, 

?)  (3) 

where  k  =  2?r/X  (X  is  the  wavelength  of  the  sound  in  air) ;  and  TO  is  the  effective  mass  per 
unit  area  of  the  wall,  which  is  of  the  order  of  0.2  to  1  times  the  actual  mass  per  unit  area. 
The  mass  reaction  of  rigid  walls  is  thus  the  dominant  factor  affecting  sound  transmission; 
the  sound  is  transmitted  largely  by  the  diaphragmlike  vibration  of  the  wall,  the  wall  re- 
sponding to  the  alternating  force  of  the  impinging  sound  wave  much  as  a  rigid  piston  would. 
In  thin  flexible  panels,  the  stiffness,  the  internal  damping,  the  size  of  the  panel,  and  the 
manner  in  which  it  is  clamped  around  the  edges  all  contribute  to  the  total  amount  of  vibra- 
tion which  will  be  imparted  to  the  partition,  and  therefore  all  these  factors  contribute  to 
the  insulation  value  of  such  walls.  However,  these  factors  are  effective  principally  at  low 
frequencies;  the  mass  is  the  predominant  factor  throughout  most  of  the  audio-frequency 


12-64 


ACOUSTICS 


range  in  determining  the  insulation  value  of  nearly  all  rigid  panels  and  partitions  encoun- 
tered in  practice.  In  Fig.  7  is  given  a  summary  of  the  measured  insulation  value  of  many 
rigid  panels  having  weights  varying  from  2  Ib  per  sq  ft  up  to  100  Ib  per  sq  ft.  There  is  a 
nearly  linear  relation  between  the  average  T.L.  and  the  logarithm  of  the  weight  per  square 
foot  of  the  partition.  (The  average  T.L.  is  the  arithmetical  mean  of  the  measured  trans- 
mission losses  at  128,  256,  512,  1024,  and  2048  cycles.) 

Whereas  the  insulation  value  of  porous  materials  is  proportional  approximately  to  the 
thickness  of  the  material,  the  insulation  value  of  a  rigid  material  increases  only  as  the 
logarithm  of  the  thickness.  Because  of  the  slow  increase  in  insulation  with  increased  mass 
or  thickness  of  a  rigid  partition,  it  is  not  always  feasible  to  secure  a  high  insulation  by 
merely  increasing  the  thickness  of  the  wall.  Thus,  it  would  be  necessary  to  increase  the 
thickness  of  a  concrete  wall  to  nearly  4  ft  hi  order  to  give  the  wall  an  insulation  of  60 jib. 


60 
55 

35° 

|    45 

i  4° 

1  35 
30 
25 
20 

A  Bureau  of  Standards 
0  Knudsen 
.  «  Meyer 
X  Nat.  Phys.  Lab. 

A    ° 

o 

X 

^ 

**** 

A 

" 

0 

-"""on 

a 
o 

0 

** 

^****° 

a 

o  A 

^ 

+** 

** 

^^** 

^-^ 

^^* 

o 

--^ 

2            3       4      5     6    7  8  9  10                  20          30     40     50   607080901 

Weight  per  Square  Foot 
FIG.  7.     Transmission  Loss  in.  db  for  Rigid  Single  Partitions 

When  walls  of  high  insulation  are  required,  it  is  more  feasible  and-  economical  to  employ 
special  structures  which  combine  the  two  principles  of  sound  insulation  just  described: 
namely,  absorption  losses  in  porous-flexible  materials,  and  inertial  losses  in  massive  parti- 
tions. Thus,  two  or  three  rigid  and  relatively  thin  partitions  separated  from  each  other 
by  felts  or  blankets  can  easily  be  composed- in  such  a  manner  as  to  give  an  insulation  of  60 
or  even  70  db.  The  insulation  value  of  many  special  forms  of  construction  employing 
these  and  other  principles  will  be  found  in  the  tables  in  the  next  section.  (See  also  ref- 
erences given  on  p.  12-75.) 

28.  COEFFICIENTS  OF  SOUND  TRANSMISSION 

The  coefficient  of  sound  transmission  of  a  panel  is  the  ratio  of  the  transmitted  to  the 
incident  sound  energy.  One  of  the  most  satisfactory  methods  for  measuring  the  coeffi- 
cients of  different  materials  is  to  make  measurements  of  the  sound  intensity  on  both  sides 
of  a  test  panel  placed  between  two  rooms  which  are  so  constructed  that  no  sound  is  trans- 
mitted from  one  room  to  the  other  except  through  the  test  panel.  Thus,  suppose  the  aver- 
age intensity  near  the  panel  in  the  source  room  to  be  1 1  and  the  average  intensity  hi  the 
test  room  to  be  Ii.  Then  the  rate  of  flow  of  sound  energy  against  the  test  panel  in  the 
source  room  will  be  I\A,  where  A  is  the  area  of  the  test  panel.  The  rate  at  which  energy 
will  be  transmitted  through  the  panel  into  the  test  room  will  be  I\Ar,  where  r  is  the  trans- 
mission _  coefficient  for  the  panel.  The  product  IiAr  is  then  the  rate  of  emission  of  sound 
energy  in  the  test  room.  Hence,  when  equilibrium  is  established,  this  rate  of  emission  of 
sound  energy  will  be  equal-  to  the  rate  of  absorption  of  sound  in  the  test  room,  which  ia 
equal  to  I^a,  where  az  is  the  total  absorption  of  the  test  room.  Hence, 


IiA 


(4) 


The  coefficient  of  transmission  thus  involves  not  only  the  intensities  on  the  two  sides  of 
tfee  panel  but  also  the  area  of  the  panel  and  the  total  amount  of  absorption  m  the  test 
room, 


COEFFICIENTS  OF  SOUND  TRANSMISSION 


12-65 


Table  2.    Coefficients  of  Sound  Transmission 


Description  of  Panel 
(Bold-face  type  signifies 
that  the  panel  has 
practical  merit) 

Weight, 
Ibper 
sq  ft 

Reduction  Factors  in  Decibels 

Au- 
thor- 
ity 

Prob- 
able 

Aver- 
age 
T.L., 
db 

Probabte 
Average 
Value 
of  r 

128 

256 

512 

1024 

2048 

Porous-flexible  Materials  and  Fiber  Boards 


Celotex,  Standard,  0.5  in.. 

0.66 

22.4 

17  3 

23  4 

27  4 

1  * 

20 

0  010 

Insulite,  0.5  in  
Hair  felt,  1  in  

0.75 
0.75 

4  9 

22.2 
4  6 

20.2 
6  0 

24.1 
7  I 

20.9 
6  7 

I* 

3 

19 

0.013 

Hair  felt,  4  in  

7.5 

12.5 

15  3 

19  7 

19  4 

3 

Rock  Wool  blanket,  0.5  in. 
covered  on  both  sides 
with  heavy  brown  paper 
TJpson  Blue  Stripe  Insula- 
tion   

15.5 
14.0 

15.1 

17.8 
16.0 

18.5 

18.4 
21.  1 

2 
2 

16 
16 

0.025 
0.025 

Thin  Rigid  Materials 


Aluminum,  0.025  in.  
Duralumin,  0.020  in  
Iron,  0.03  in.  galvanized  .  . 
Lead,  0.062  in  

0.35 
0.33 
1.2 
3  9 

17.9 
14.1 
25.3 
31  ft 

13.2 
12.5 
20.5 
33  2 

17.7 
17.6 
28.8 
32  0 

23.2 
22.5 
35.0 
32  I 

I  * 
1  * 
1  * 
1  * 

16 
15 
25 
30 

0.025 
0.032 
0.0032 
0  0010 

Plywood,  0.25  in.,  three- 
ply  

0.73 

21.0 

20  7 

25  5 

26  0 

I  * 

21 

0  OO&G 

Mahogany,  1.85  in  

4.9 

26.0 

27  0 

36  0 

4f 

28 

0.0016 

Plaster  board,  0.5  in  

27.0 

28.0 

33.0 

28 

0.0016 

Doors  and  Windows 


Doors 

Birch  veneer,   light,  four 
panel.  .  .  

13.0 

16  1 

20  4 

22.8 

22.0 

3 

22 

0.0063 

"Cold-storage"  door, 
double  wall,  4  in.  ...... 

16.4 

20.8 

27.1 

29.4 

28.9 

3 

29 

0.0013 

Oak,  solid,   1175  in.,  with 
cracks  as  ordinarily  hung 

11.5 

15.1 

20.4 

22.0 

16.2 

3 

20 

0.01 

Oak,  like  above,  well  seas- 
oned and  air  tight  

15.1 

18.2 

22.8 

25.7 

25.2 

3 

25 

0.0032 

Steel,  solid,  0.25  in.  

25.1 

26.7 

31.1 

36.4 

31.5 

3 

35 

0.00032 

Window 

T5 

Glass,  plate,  0.25  in  
Glass,  plate,  0.25  in.,  four 
panes 

3.5 

23.2 

32.6 
20.8 

30.9 
26.4 

33.5 
27.5 

34.2 
22.8 

1* 
3 

30 
29 

0.0010 
0.0013 

Glass,     plate,     0.25    in., 
double    glazed,     16    in. 
separation  

43.  Ql 

3 

48 

0.000016 

Rigid  Partitions  (Tile,  Brick,  Concrete,  etc.) 


Brick  panel,  8  in.;  plas- 
tered both  sides  

97.0 

47.7 

49.4 

57.0 

59.2 

1* 

49 

0.500013 

Tiles,    hollow-clay    parti- 

tion, three  cells,  4  in.  by 

12  in.  by   12  in.,  plas- 
tered both  sides  

29.0 

41.1 

40.0 

41.5 

49.9 

1  * 

40 

0.00010 

Tile,   hollow  clay,    4  in., 
unplastered        ....... 

17.0 

24.5 

24.1 

26.1 

35.5 

29.8 

3 

35 

0.00032 

Tile,   like  above,    0.5  in. 

plaster     .     ........... 

22.0 

25.1 

24.3 

26.9 

38.2 

33.9 

3 

38 

0.00016 

Tile,  hollow  gypsum,  3  in,, 
unplastered              ..... 

11.  1 

19.2 

18.7 

20.8 

28.5 

30.0 

3 

31 

0.00080 

Brick  4.5  in.,  0.5  in.  fiber 

board  stuck  on  each  face, 

then  0.6  plaster  on  each 
faee                     

46.8 

44. 

45. 

51. 

73. 

6 

48 

0.000016 

Brick,  0.  25  in.  Masonite  on 
lath  on  one  side,  11  in.  , 

88.0 

36.5 

37.0 

48.0 

58.5 

61.0 

5§ 

48 

0.000016 

12-66 


ACOUSTICS 


Table  2.    Coefficients  of  Sound  Transmission — Continued 


Description  of  Panel 
(Bold-face  type  signifies 
that  the  panel  has 
practical  merit) 

Weight, 
Ibper 
sqft 

Reduction  Factors  in  Decibels 

Au- 
thor- 
ity 

Prob- 
able 
Aver- 
age 
T.L., 
db 

Probable 
Average 
Value 
of  T 

128 

256 

512 

1024 

2048 

Rigid  Partitions  (Tile,  Brick,  Concrete,  etc.)—  Continued 

Clinker    concrete,    3    in., 
plastered  both  sides  
Concrete,   3  in.;  0.75  in. 
cement  and  linoleum  .  .  . 

31. 

49.0 

28. 
36.5 

33. 
37.5 

40. 
44.5 

50. 
54.0 

57. 
65.0 

6 

5§ 

40 
48 

0.00010 
0.000016 

Wood  Studs  and  Plaster,  Metal  Channel  Iron  and  Plaster,  etc. 

Wood  studs,  2  in.  by  4  in., 
17in.o.c.,0.25in.by  1.5 
in.  wood  lath,  0.375  in. 
apart,  gypsum  scratch, 
lime  brown,  smooth  fin- 
ish 

49.5 

42.6 

32.2 
46.6 

43.7 

52.2 

35.7 
42.9 

36.7 

1  * 

3 
3 

3 

44 

37 
43 

40 

0.000040 

0.00020 
0.000050 

0.00010 

Wood   stud;    wood   lath, 
0.25  in.  by  1.  5  in.  spaced 
0.375   in.;    gypsum 
scratch   coat,    0.25  in.; 
brown   coat,    0.25   in.— 
0.375  in.;  finish  coat  
Wood  studs,  etc.,  as  above 
except  lime  plaster 

18.6 
18.0 

12.0 

24.4 
27.5 

17.7 

25.6 
28.8 

24.7 

29.1 
38.1 

37.0 

Wood  studs,  2  in.  by  4  in., 
0.5  in.  Celotex,  gypsum 
plaster  

Floor  and  Ceiling  Partitions 


Concrete  flat  slab  floor 
construction,  reinforced. 
Insulite  furred  out,  ap- 
plied as  ceiling 

54.4 

50.9 

54.8 

58.7 

56.5 

53.2 

1  * 

51 

0.0000080 

Wood  joists.  Lower  side 
plastered  on  wood  lath; 
upper  side,  subflooring 
and  0.375  in.  finish  floor- 

47.9 

46.8 

40.7 

50.1 

48.8 

1  * 

43 

0.000050 

Wood  joists,  etc.,  as  above, 
with  floating  floor  con- 
sisting of  nailing  strips 
rough  and  finish  flooring 

57.6 

57.5 

54.8 

62.4 

57.6 

1  * 

53 

0.000005 

Double  Walls 


Tile,  double  2  in.  solid  gyp- 

sum,   unplastered,    un- 

bridged,  2  in.  separation; 

structurally  separated  .  . 

20.4 

25.2 

34.2 

44.5 

51.0 

62.6 

3 

48 

0.000016 

Tile,  etc.,  as  above,  except 

bridged,  at  middle  

20.4 

21.3 

32.7 

37.0 

45.6 

52.0 

3 

44 

0.000040 

Tile,  etc.,  as  above,  filled 

with  sawdust  

23.0 

21.6 

28.1 

39.3 

47.0 

54.0 

3 

44 

0.000040 

Tile,  double  2  in.  solid  gyp- 

sum,   unplastered,    un- 

bridged,  4  in.  separation; 

structurally  separated  .  . 

20.4 

28.4 

47.4 

54.2 

59.0 

56.8 

3 

51 

0.000008 

*  Reduction  factors  from  the  Bureau  of  Standards,  for  the  most  part,  are  for  the  following  frequency 
bands:  150  to  157,  250  to  285,  500  to  547,  1000  to  1070,  and  20OO  to  2175  cycles.  The  data  for  the 
several  frequency  bands  are  recorded  under  the  frequencies  to  which  the  frequency  bands  most  nearly 
correspond. 

t  For  frequencies  of  300,  500,  and  1000  cycles. 

t  Average  value  from  128  to  2048  cycles. 

§  Berg  and  Holtsmark's  reduction  factors  are  for  frequency  bands  of  100-200,  200-400,  400-800, 
800-1600,  and  1600-3200  cycles.  Their  values  are  recorded  under  128,  256,  512,  1024,  and  2048  cycles, 
respectively. 

Authority:  (1)  Bureau  of  Standards,  (2)  V.  O.  Knudsen,  (3)  P.  E.  Sabine,  (4)  Davis  and  Littler, 
<5)  Berg  and  Holtsmark,  and  (6)  National  Physical  Laboratory. 


CONSIDERATIONS  IN  SELECTION   OF  MATERIALS      12-67 

For  other  methods  of  measuring  transmission  coefficients  see  Knudsen,  Architectural 
Acoustics,  Chapter  XI. 

The  results  of  sound-insulation  measurements  obtained  in  different  laboratories  axe 
grouped  in  Table  2.  Each  group  contains  the  data  for  materials  or  partitions  having  a 
number  of  properties  in  common.  The  data  in  each  group  have  been  arranged  so  as  to 
keep  together  those  from  each  laboratory,  which  is  necessary  since  the  results  on  the  same 
panel  tested  in  different  laboratories  are  not  always  in  good  agreement.  The  data  from 
most  laboratories  give  what  has  been  called  the  reduction  factor  for  the  panel  or  partition 
tested.  This  reduction  factor  is  usually  the  ratio  of  the  intensities  of  sound  on  both  sides 
of ^  the  test  panel,  or,  in  decibels,  is  10  times  the  logarithm  of  this  ratio.  Since  the  trans- 
mission coefficient  depends  upon  the  size  of  the  panel  and  the  amount  of  absorption  in 
the  test  room,  as  is  shown  by  eq.  (4),  the  reduction  factors  published  by  several  labora- 
tories (before  about  1935)  do  not  agree  with  the  T.L.,  which  in  decibels  is  10  logie  1/r.  The 
compiler  has  made  an  attempt  to  adjust  the  data  from  different  laboratories  in  such  a 
manner  as  to  give  comparable  ratings  for  all  the  materials  and  partitions  listed  in  the  tables. 

In  the  table,  the  data  given  under  "Reduction  Factors  in  Decibels"  are  the  results 
published  by  the  authors.  The  data  given  in  the  last  two  columns  in  the  table  give  the 
compiler's  estimate  of  the  probable  average  value  of  the  T.L.,  and  the  probable  average 
value  of  the  transmission  coefficient  r.  The  reduction  factors  at  the  different  frequencies 
are  useful  since  they  describe  the  insulation  value  of  the  panel  or  partition  at  these  fre- 
quencies. This  is  often  an  important  matter  in  the  selection  of  materials  or  partitions 
for  sound  insulation.  For  example,  partitions  which  have  a  relatively  low  insulation 
value  in  the  frequency  range  from  500  to  1000  cycles  would  not  be  suitable  for  the  insula- 
tion of  traffic  noise  and  of  most  noises  met  in  buildings,  since  most  such  noises  contain  a 
relatively  large  amount  of  sound  energy  in  this  frequency  range.  (See  Fig.  1.)  For  this 
reason,  the  panels  which  show  the  highest  values  of  T.L.  may  not  supply  the  greatest 
amount  of  sound  insulation  for  all  types  of  noise.  It  is  necessary,  in  determining  the  best 
type  of  partition  for  each  problem  which  arises  in  sound  insulation,  to  give  consideration 
to  the  reduction  factors  at  the  different  frequencies  as  well  as  to  the  average  value  of  T.L. 
or  r. 

Partitions  which  possess  outstanding  merit  with  regard  to  both  insulation  value  and 
practicability  are  designated  by  bold-face  type  in  the  column  which  gives  the  name  and 
description  of  the  partition,* 

29.  PRACTICAL  CONSIDERATIONS  IN  THE  SELECTION  OF 

MATERIALS  AND  TYPES  OF  STRUCTURE  FOR 

INSULATION  IN  BUILDINGS 

Since  errors  as  large  as  3  or  4  db  are  inherent  in  many  of  the  data  on  sound  insulation, 
small  differences  in  the  tabulated  results  of  the  preceding  article  should  not  be  regarded 
as  having  much  significance.  A  consideration  of  the  tabulated  data  suggests  the  following 
generalizations  concerning  the  insulative  properties  of  different  building  materials  and 
partitions : 

1.  The  insulation  value  of  rigid  masonry  or  monolithic  partitions  increases  directly  as 
the  logarithm  of  the  weight  per  square  foot  of  wall  section — so  that  the  rate  of  increase  of 
insulation  with  increased  weight  is  relatively  slow  for  partitions  which  are  heavier  than, 
say,  30  or  40  Ib  per  sq  ft.    As  a  consequence,  it  is  often  in  the  interest  of  both  insulation 
and  economy  to  substitute  two  or  more  light-weight  partitions,  or  specially  composed 
partitions,  for  heavy  masonry  partitions.    There  are  many  occasions  in  building  practice 
where  this  may  be  done  with  a  gain  in  insulation,  with,  a  reduction  in  the  dead  load  of  the 
building,  and  at  a  reduced  cost.    However,  for  thin  partitions,  where  the  dead  load  is  not 
a  serious  problem,  dense  rigid  panels  such  as  plastered  brick  or  solid  tile  provide  a  satis- 
factory means  of  obtaining  a  T.L.  of  about  40  to  45  db. 

2.  Lime  plaster  on  wood  lath  and  wood  studs  gives  better  insulation  than  an  equal 
thickness  of  gypsum  plaster  on  wood  lath  a-rtrl  wood  studs.    Sabine  reports  an  advantage 
of  about  9  db  for  lime  plaster  as  compared  with  gypsum  plaster,  and  the  Bureau  of  Stand- 
ards reports  an  even  greater  difference.    Wood  stud  and  plaster  partitions  rate  slightly 
higher  than  channel  iron  and  plaster. 

3.  Wood   partitions*   with  tongue-and-groove  joints,   provide  more  insulation   than 
masonry  partitions  of  the  same  weight  or  thickness.    (The  development  of  cracks  in  wood 
partitions,  however,  will  greatly  reduce  their  insulation  value,) 

*  More  extensive  tables  on  sound-insulation  data  will  be  found  in  Knudsen's  Architectural  Acoustic** 
Glover's  Practical  Acoustics  for  the  Constructor,  Constable  and  Ashton,  Sound  Insulation  of  Single  and 
Complex  Partitions,  Philosophical  Mag.,  Vol.  23,  161-181  (1937) ,  and  Knudsen  and  Harris,  Acoustical 
Designing  in  Architecture  (1949). 


12-68 


ACOUSTICS 


4.  Double  partitions  seem  to  offer  the  most  feasible  means  of  obtaining  high  insulation 
at  a  reasonable  cost  and  reasonable  dead  load.  The  separate  partitions  should  be  as  com- 
pletely insulated  from  each  other  as  possible,  as  the  introduction  of  structural  ties  between 
the  separate  partitions  tends  to  convert  the  two  partitions  into  a  single  rigid  one  and  thus 
greatly  reduces  the  insulation.  The  suspension  of  a  blanket  or  fiber  board  between  double 
partitions  or  between  the  wood  studs  or  channel  irons  in  staggered  stud  partitions  is  often 


•8  Brick 


I 

m\     T.L-49d6 
H— 3wHoIIow  Clay  Tile 


T.L.=40dd 


-  4HoHow  Clay  Tile 
""l"x2"Furring  Strips 
'Paper  and  Metal  Lath 
"Gypsum  Plaster 
T.L=52<Z6 


-%  Steel  Channels 
-^'Plaster 

Gypsum:  T.  L»34dZ> 
Lime  .•  T.  L.  =40  db  • 


Gypsum  Scratch  Coat 
-*"  Lime  Brown  and  Finish 
—Wood  Lath 

\  2\  4"Wood  Studs 


-2"x  4*Wd  Stud* 
-Sheet  Metal 
*-3  Coat  Gypsum 


T.L-45d6 


ough  aiid  Finished  Flooring 
ood  Nailing  Strips 
Concrete  Slab 

re  Board  and  Plaster 


-Rough  and  Finished  Flooring 
•Wood  Nailing  Strips 
"Hollow  Clay  Tile 
Vibre  Board  and  Plaster 


.Floated  Rough  and  Finished 
Flooring 

2x  10  Wood  Joists 
Plaster  on  Wood  Lath 


-4*HoHow  Clay  Tile 
-l"x  2"  Furring  Strips 
'1$  Fibre  Board 
-Gypsum  Plaster 

T.  L— 54  db 


.2x4  Wood  Studs 

.  t  Fibre  Board, 
Joints  Filled 
}  Coat  Gypsum 

T.L=49d6 


.-Rough  and  Finished  Flooring 
-Absorptive  Blanket 

Plaster  on  Wood  Lath 
T.  L.>50  db     Good  Insulation  for 
Fool  Falls,  etc, 


-3  Solid  Gypsum  Tile 
"  Plaster 


-4  Brick 
-%*PIast« 


Staggered  Wood  Studs 
-Absorptive  Blanket 
-Plaster  on  Fibre  Board 

T,L>50d6 

2  Layers  J/Mutex  with 
Sheet  of  28  Gauge 

Iron  between 
—3  Masonry 
-*-  Plaster 


T.  L>60  db 


T,  L  -47  db  JUfn  Acous.  Corp.  of  America 

Double  Masonry  Wall 
T.  L=>60  db 
FIG.  8.    Recommended  Types  of  Structure  for  Sound  Insulation 


Rough  and  Finished  Flooring 
Ji' Fibre  Board 
oncrete  Slab 
Hung  Ceiling 


Flexible  and  Absorptive 

Material 
Resilient  Chair 
Concrete  Slab 
Resilient  Hanger 
Plaster  and  Lath 
U.  S.  Gypsum  and 
Johns-Manville 
System 


an  aid  to  insulation.  The  addition  of  any  absorptive  material  in  the  air  space  between 
double  partitions  contributes  considerably  to  sound  insulation  unless  the  absorptive 
material  makes  a  rigid  or  semirigid  bridge  between  the  two  partitions,  hi  which  case  it 
may  be  worse  than  nothing.  Thus,  the  addition  of  cinders,  pumice,  or  other  rigid-porous 
materials  between  structurally  separated  partitions  will  sometimes  reduce  the  overall 
insulation.  * 

A  number  of  satisfactory  types  of  construction  for  obtaining  wall  partitions  having  a 
T.L.  greater  than  40  db  and  floor  and  ceiling  partitions  having  a  T.L.  greater  than  50  db 
are  indicated  in  Fig.  8.  These  methods  of  construction  will  meet  most  of  the  requirements 
lor  sound  insulation  that  will  arise  in  connection  with  the  design  of  buildings.  The  aver- 
age TX.  is  given  for  each  partition.  (See  also  references  given  on  pp.  12-67  and  12-75.) 


THE  HEARING  OF  SPEECH  IN  AUDITORIUMS        12-69 


30.  CALCULATION  OF  INSULATION  IN  BUILDING  DESIGN 

From  a  simple  consideration  of  the  transmission  of  sound  through  the  boundaries  of  a 
room  it  can  be  shown  that  the  noise-reduction  factor  for  a  room  (the  number  of  decibels 
the  intensity  of  the  sound  is  reduced  by  transmission  through  the  boundaries)  is  given  by 


Noise-reduction  factor  =  10  logw  7 


(5) 


where  a  is  the  total  absorption  of  the  room,  and  T  —  r\A\  +  rzA*  -f-  -riA*  +  .  . .  is  the 
total  transmittance  for  th,e  boundaries  of  the  room.  TI,  TS,  and  ra  are  the  coefficients  of 
transmission  of  the  different  parts  of  the  boundaries  of  the  room,  and  J.i,  A^  and  A*  are 
the  corresponding  areas  of  these  boundaries. 

In  order  to  illustrate  the  use  of  eq.  (5)  a  typical  calculation  will  be  made  for  determin- 
ing the  noise-reduction  factor  of  a  small  studio. 
Volume  of  room  =  50,000  cu  ft. 
Total  absorption  in  room,  including  audience  =  2400  sabins. 

Description  of  Walls,   Ceiling,  Windows,  and  Boors;  and  the  Transmittance  through 

These  Surfaces 


Material 

Area,  At 

sqft 

T 

rA 

4-in.   concrete  slab  ceiling  plus   1/2  in.  acoustical 

2500 

0.000025 

0.0625 

8~in.  brick  walls  plus  V9  in.  acoustical  material.  ..... 

4500 

.0000080 

.0360 

3/ig-in.  glass  windows,  closed 

400 

.00110 

.440 

}  1/o-frj,  hardwood  <^f>ors,  goo*i  closure 

100 

.00031 

.031 

Total  transmittance  (T)  

0.5695 

fore                                        a      V      2400 

f0r°                                        T  ~  I,  ~  0.5695  ~  L 

and 


10  logio  4210  =  36.2  db 


That  is,  the  noise-reduction  factor,  or  the  effective  insulation  which  the  room  provides 
against  outside  noise,  is  36.2  db.  Thus,  if  the  studio  is  located  where  the  outside  noise  is 
at  a  level  of  60  db,  the  level  of  the  noise  which  reaches  the  studio  will  be  60  —  36.2  or 
23.8  db.  It  will  be  noted  that  most  of  the  transmitted  noise  is  that  which  comes  through 
the  glass  windows,  and  that  therefore  the  noise-reduction  factor  can  be  increased  con- 
siderably by  means  of  double  windows  or  by  dispensing  with  the  windows. 


ACOUSTIC  DESIGN  OF  AUDITORIUMS 

By  Vent  O.  Knudsen 

31.  THE  HEARING  OF  SPEECH  IN  AUDITORIUMS 

Four  principal  factors  affect  the  hearing  of  speech  in  auditoriums:  the  shape  of  the  room, 
the  loudness  of  the  speech  which  reaches  the  listeners,  the  reverberation  characteristics 
of  the  room,  and  the  amount  of  noise  in  the  room.  If  average  speech  is  loud  and  distinct, 
and  entirely  free  from  the  interfering  effects  of  noise  and  reverberation,  the  percentage 
articulation  for  the  average  listener  will  be  96,  that  is,  96  out  of  100  meaningless  monosyl- 
labic speech  sounds  will  be  heard  correctly. 

In  an  equation  for  calculating  the  percentage  articulation  for  a  room  it  will  be  necessary 
to  introduce  reduction  or  distortion  factors  due  to  (1)  the  shape  of  the  room,  (2)  inadequate 
loudness,  (3)  excessive  reverberation,  and  (4)  extraneous  noise.  It  is  possible  to  represent 
approximately  the  percentage  articulation  in  any  room  by  the  equation 

Percentage  articulation  —  QGkJciktk*  (1) 

where  k,  —  the  reduction  factor  due  to  the  shape  of  the  room. 

ki  —  the  reduction  factor  due  to  the  inadequate  loudness  of  speech. 

AT  =  the  reduction  factor  due  to  the  excess  of  reverberation  in  the  room. 

k^  =  the  reduction  factor  due  to  the  extraneous  noise  in  the  room. 


12-70 


ACOUSTICS 


In  the  ideal  room  each  of  these  factors  will  be  equal  to  unity,  so  that  the  percentage 
articulation  under  such  conditions  would  be  96  per  cent. 

Experimental  data  have  been  obtained  by  means  of  which  it  is  possible  to  determine  the 
appropriate  values  of  these  four  factors  for  any  auditorium,  although  considerable  re- 
search remains  to  be  done  before  accurate  values  of  ks  are  known.  When  these  factors 
are  determined  for  a  certain  auditorium  and  substituted  in  eq.  (1)  the  resulting  product 
gives  the  probable  average  percentage  articulation  in  that  auditorium. 

It  is  admittedly  only  an  approximation  to  represent  these  factors  by  single  numbers. 
For  example,  the  reverberation  is  described  not  by  a  single  number  but  by  a  curve  giving 
the  times  of  reverberation  at  different  frequencies.  However,  if  one  takes  the  time^  of 
reverberation  for  a  tone  of  512  cycles,  one  will  have  a  fairly  reliable  index  for  representing 
the  condition  of  reverberation  in  a  room,  especially  if  the  reverberation  time  does  not  vary 
too  widely  at  different  frequencies.  Such  a  single  index  is  very  useful  for  rating  the 
acoustical  quality  of  rooms.  In  problems  of  design,  on  the  other  hand,  consideration 
should  be  given  to  the  reverberation  throughout  the  entire  range  of  frequencies.  Similarly, 
the  noise  spectrum  in  the  room  and  the  shape  of  the  room  cannot  be  represented  rigorously 
by  single  numbers.  However,  in  the  rectangular  room  of  conventional  shape  it  is  not  only 
admissible  to  use  a  single  factor  to  represent  the  effect  of  shape  on  articulation,  but  the 
factor  ka  will  not  deviate  appreciably  from  unity. 

The  four  factors  which  affect  the  hearing  of  speech  in  rooms  will  now  be  considered 
separately. 

SHAPE.  It  is  not  only  necessary  to  avoid  shapes  which  will  produce  acoustical  defects, 
such  as  echoes,  flutters,  sound  foci,  interfering  effects,  and  "whispering  gallery"  effects, 
but  it  is  of  prime  importance  to  design  shapes  which  will  facilitate  the  most  advantageous 
flow  of  sound  energy  to  all  auditors  in  the  room. 

There  axe  three  outstanding  forms  which  should  never  be  tolerated:  (1)  those  which  will 
produce  a  pronounced  focusing  of  sound,  thus  giving  an  excessive  concentration  of  sound 
in  some  places  and  a  scarcity  of  sound  in  other  places;  (2)  those  which  will  produce  exces- 
sive delays  between  the  sound  which  reaches  the  auditors  by  a  direct  path  from  the  source 
a-nd  that  which  reaches  the  auditors  by  reflection  from  the  ceiling  or  walls ;  and  (3)  those 

in  which  the  sound  reaching  the  auditors 
travels  a  relatively  long  distance,  at 
grazing  incidence,  over  a  highly  absorp- 
tive surface.  The  sound  which  comes 
by  the  reflected  paths  always  has  to 
travel  a  greater  distance  than  that  which 
comes  by  the  direct  path,  and  if  the 
difference  in  these  path  lengths  is  as 
great  as  65  ft  the  reflected  sound  will  be 
delayed  to  the  extent  that  it  is  heard  as 
a  separate  sound;  that  is,  the  delayed 
sound  produces  an  echo.  Even  when 
the  reflected  sound  is  delayed  as  much 
as  50  ft  it  unites  with  the  direct  sound 
sufficiently  out  of  phase  to  produce  a 


FIG.  1.     Kefleetion  of  Sound  from  a.  Domed  Ceiling 


masking  or  blurring  interference.  Figure  1  exhibits  a  characteristic  defect  which  results 
from  the  use  of  concave  surfaces. 

Figure  2  shows  a  longitudinal  section  of  a  cut-away  model  of  an  auditorium  which  is 
not  only  free  from  concentrations,  dead  spots,  and  interfering  reflections  but  also  is  so 
shaped  as  to  give  a  nearly  uniform  distribution  of  diffusely  reflected  sound  to  all  parts  of 
the  auditorium,  with  a  slight  preference  for  the  more  remote  parts. 

In  many  auditoriums,  and  even  in  some  sound-recording  studios,  it  may  be  difficult 
or  even  impossible  to  avoid  large  and  therefore  troublesome  differences  of  path  between 
the  direct  and  reflected  sound.  In  such  instances  it  is  advisable  to  break  up  the  surfaces 
producing  these  delayed  reflections  by  introducing  coffers,  beams,  pilasters,  or  other  irregu- 
larities in  contour.  A  number  of  rooms,  highly  acclaimed  for  their  good  acoustics,  have 
been  designed  with  walls  and  ceiling  deliberately  covered  with  polycylindrical  sound 
diffusers.  Figure  3  is  a  typical  example.  In  rooms  where  public-address  systems  are 
required,  the  architect  will  have  greater  freedom  in  designing  the  shape  and  size  of  the 
room. 

Both  speech  and  music  rooms  should  be  designed  so  that  the  auditors  receive  a  relatively 
large  amount  of  sound  which  travels  directly  from  the  source  or  from  reflectors  located 
sufficiently  near  the  source  so  that  this  reflected  sound  is  nearly  in  phase  with  the  direct 
sound.  The  stage,  pulpit,  platform,  choir  loft,  or  other  location  for  the  speaker  or  per- 
former should  be  well  elevated  above  the  audience  and  provided  with  large,  reflective 


1.  Acoustically    treated    projection    booth  with  sound 
amplification   equipment  and   controls    for  sound 
monitoring. 

2.  Ceiling    planes  reflect  sound   to  all   parts  of  the 
auditorium. 

3.  Three-channel  public  address  system  to  reproduce 
stage  sound  in  "auditory  perspective". 

4.  High-fidelity  speakers:  bass-compensated  dynamic 
speaker  for  low  tones;   high  frequency  directional 
horns  for  high  tones. 

5.  Backstage  treated  with  acoustical  plaster  to  reduce 
"stage  echoes". 

6.  Acoustic  treatment  on  walls:  over-all  distribution 
in  alternate  bands     of  (a)  acoustic    tile  and  (6) 
hacdwall  plaster, 

FIG.  2.     Cut-away  Model  of  the  High  School  Auditorium  for  "Whittler,  California.    (William  Harrison, 

Architect.) 


7.  Proscenium  splays,  horn-like  shape  of  stage  opening 
projects  sound  to  audience. 

8.  Upholstered   seats:   absorption    value  of  each  seat 
equivalent  to  that  of  a  person's  clothing. 

a.  Double  doors   to   foyer  insulate  against  external 

noises. 
10.  Slanting  rearwalls  on  main  floor  and  balcony  reflect 

sound  down  toward  rear  seats. 

It.  Acoustically  treated  foyer  to  reduce  external  noises. 
12,  Streamlined  balcony    improves  flow    of  sound  to 

rear  seats. 


FIG  3,     Polyeylindrical  Sound  Diffusers  in  an  RCA  Disk  Recording  Studio  in  New  York.    (Volkmann.) 

12-71 


12-72 


ACOUSTICS 


surfaces  located  behind,  above,  and  on  both  sides  of  the  position  where  the  sound  °rigmates. 
In  addition,  especially  in  large  rooms,  the  floor  should  rise  progressively  toward  the  rear 
of  the  room,  or  the  room  should  be  designed  with  one  or  more  balconies  so  that  all  auditors 
obtain  an  abundance  of  direct  or  beneficially  reflected  sound-the  path  length  o .the  re- 
flected sound  should  not  exceed  that  of  the  direct  sound  by  more  than  50  ft,  and  Preferably 
less  than  that,  so  that  the  reflected  sound  will  reinforce,  and  not  interfere  with,  the  direct 

sound.      Long  rooms   with   level 


110 


lbel 

8 


tonslty  Uvtfl 

8  8 


Room  B  T70f  x  200' x; 
Absorbent  Ceiling 


Room  A  360'  x1  560'  x  27' 
Non-absorbent  -Celling 


200 


floors,  with  a  low  platform  or  no 
platform  at  all,  and  with  a  highly 
absorptive  ceiling,  are  especially 
poor  speech  rooms,  since  much 
of  the  sound  reaching  the  auditors 
in  such  rooms  travels  at  near 
grazing  incidence  over  the  highly 
absorptive  audience  and  also  along 
the  absorptive  ceiling.  (See  Fig. 
4.)  The  intensity  of  free  progres- 
sive sound  waves  propagated  over 
such  surfaces  has  been  found  to 
diminish  as  the  inverse  fourth 
power  of  the  distance  from  the 
sound  source.  Properly  designed 


0  50  100  150 

Distance  from  the  Source  In  Feet 

FIG.  4.     Variation  of  Intensity  of  Sound  with  Distance  from    . 

Source  in  Large  Rooms  with  Absorbent  and  Non-absorbent  reflective  surfaces,  large  compared 
Ceilings,  Showing  the  Excessive  Decrease  in  Level  Which  Oc-  -+-L  +T^  TTTOTrola-no-f-'h  r>f  +h<*  «innnr? 
curs  When  Sound  Travels  over  or  along  an  Absorptive  Surface  W1jf  ltne  waveiengi:n  01  une  bouna, 

will  largely  compensate  for  such 

excessive  losses  and  will  insure  adequate  loudness  of  unamplified  speech  for  rooms  having 
volumes  of  less  than  about  50,000  cu  ft.  For  larger  rooms  it  is  necessary  to  amplify  the 
speech  with  a  suitable  public-address  system. 

For  further  details  regarding  shape  of  rooms  consult  Bagenal  and  Wood,  Planning  for 
Good  Acoustics;  Knudsen  and  Harris,  Acoustical  Designing  in  Architecture;  and  the  current 
and  bound  volumes  of  the  Journal  of  the  Acoustical  Society  of  America. 

NOISE.  The  curve  in  Fig.  5,  obtained  empirically,  gives  the  value  of  the  noise- 
reduction  factor  kn  for  different  amounts  of  noise.  The  abscissa  is  the  ratio  of  the  sound 
level  of  the  noise,  in  decibels,  to  that  of  the  speech,  also  in  decibels.  Thus,  when  the  noise 

1.0 


.8 


.6 
•as 

.4 


.2  .4  .6  .8 

Ratio  of  Noise  to  Speech  Levels 
FIG.  5.     Effect  of  Noise  on  the  Hearing  of  Speech 


1.0 


is  at  the  same  level  as  the  speech,  the  abscissa  is  1.0.  Although  the  value  of  kn  given  in  these 
curves  is  only  an  approximation,  based  upon  a  limited  number  of  measurements,  experi- 
ence has  indicated  that  it  is  useful  in  problems  of  design  or  correction. 

LOUDNESS.  The  curve  of  Fig.  3,  p.  12-31,  gives  the  percentage  articulation  for 
speech  as  a  function  of  the  sound  level  of  speech,  as  determined  by  Fletcher  and  Steinberg. 
Thus,  the  optimal  sound  level  of  speech,  in  quiet  surroundings,  is  70  db  above  threshold, 
and  for  levels  below  40  db  the  articulation  drops  off  rapidly.  The  loudness  reduction 


THE  HEARING  OF  SPEECH  IN  AUDITOBIUMS         12-73 


factor  ki  to  be  used  in  eq.  (1)  is  obtained  by  dividing  the  percentage  articulation  by  96. 
The  curve  in  Fig.  6  gives  the  average  speech  power,  in  microwatts,  of  the  average  speaker 
in  rooms  of  different  size.  By  means  of  this  curve,  and  the  amount  of  absorption  in  the 


100 


8 


8 


354        7Q7       14.14       2330     5660     11J30G  22,600   45,200  Gi 


U500  25,000  50,000   100,000200,000  400,000800,0001,600,000  Cn.Ft 
Volume 

FIG.  6.     Average  Speech  Power  of  Speakers  in  Rooms  of  Different  Sizes 

room,  it  is  possible  to  calculate  the  average  sound  level  of  speech  for  any  auditorium,  and 
then  from  Fig.  3,  p.  12-31,  to  calculate  the  appropriate  value  of  ki.  In  Table  1  are  given 
values  of  ki  for  rooms  of  different  size  and  different  times  of  reverberation. 

Table  1.    Values  of  ki  for  Use  in  Eq.  (1) 


Time  of 

Volume  of  Room,  cu  ft 

tion,  sec 

12,500 

25,000 

50,000 

100,000 

200,000 

400,000 

800,000 

1,600,000 

0.50 

0.96 

0.94 

0.92 

0.90 

0.88 

0.85 

0.81 

0.76 

0.75 

.97 

.95 

.93 

.91 

.89 

.87 

.84 

.80 

1.00 

.97 

.96 

.94 

.92 

.90 

.88 

.86 

.82 

1.25 

.97 

.96 

.95 

.93 

.91 

.89 

.87 

.83 

1.50 

.98 

.96 

.95 

.94 

.92 

.90 

.88 

.84 

2.00 

.98 

.97 

.96 

.95 

.93 

.91 

.89 

.86 

3.00 

.98 

.97 

.97 

.96 

.94 

.92 

.91 

.88 

4.00 

.98 

.98 

.97 

.96 

.95 

.94 

.92 

.89 

6.00 

.99 

.98 

.98 

.97                .96 

.95 

.93 

.91 

8.00 

.99 

,99 

.98 

.97       1         .96 

.95 

.94 

.92 

REVERBERATION.  The  curve  shown  in  Kg.  7,  empirically  determined,  gives  the 
value  of  kr,  the  reverberation  reduction  factor,  for  different  times  of  reverberation  at  512 
cycles.  This  curve  is  based  upon  the  results  of  speech  articulation  data  obtained  in  four- 


LO 
.8 
.6 
A 
.2 

0 

1 

—  *=< 

"-^ 

•^ 

•s,^ 

^^•v. 

^ 

"•^ 

U"""""-S—  N 

. 

—      — 

5        1.0       2.0       3.0       4.0      5.0       6.0       7.0       8.0       9. 
Time  of  Re  verfaeratkjn—  Seconds 
FIG.  7.    Effect  of  Reverberation  on  Hearing  of  Speech 

teen  different  auditoriums  having  times  of  reverberation  varying  from  less  than  1  sec  up 
to  more  than  8  sec.  It  is  seen  that  kr  decreases  almost  uniformly  as  the  time  of  reverbera- 
tion increases  from  1  to  about  6  sec.  The  abscissa  in  this  curve  gives  the  reverberation 


12-74 


ACOUSTICS 


time  at  512  cycles.  In  general,  the  time  of  reverberation  at  128  cycles  should  be  about 
25  to  50  per  cent  longer  than  it  is  at  512  cycles,  and  the  reverberation  time  should  remain 
approximately  constant  at  frequencies  above  512  cycles,  increasing  slightly  for  frequencies 
above  2048  cycles. 

COMBINED  EFFECTS  OF  LOUDNESS  AND  REVERBERATION.  Since  the  addi- 
tion of  absorption  to  a  room  diminishes  the  loudness  of  speech  as  produced  by  the  average 
speaker,  it  is  reasonable  to  assume  that  there  will  be  an  optimal  time  of  reverberation  for 
speech  rooms.  This  optimal  time  will  be  attained  when  a  further  reduction  in  the  reverber-  • 
ation  will  concurrently  reduce  the  loudness  of  speech  to  the  extent  that  the  impairment 
produced  by  the  diminished  loudness  will  just  compensate  for  the  improvement  occasioned 
by  the  reduction  of  the  reverberation.  The  curves  shown  in  Fig.  8  were  calculated  by 
means  of  eq.  (1),  using  the  appropriate  values  of  the  loudness-reduction  and  reverberation- 
reduction  factors.  These  curves  give  approximately  the  average  percentage  articulation 


1UU 

90 
c  80 

JO 

«  70 

Iso 

1 

&  50 

40 
30 

c 

,urve 
Cu.  Ft. 
(a)      25,000. 
(6)    100,000. 
(c)    400,000. 
(d)    800,000. 
»  1,600,000. 

Volume 
Cu.  M. 
707 
2,830 
11,300 
22,6CO 
45,200 

(ay— 

(vr~ 

=^ 

s^ 

& 

t~\s 

^ 

^ 

^s" 

w 

^x 

^ 

^ 

^ 

% 

^§^5»% 

^^ 

^ 

LO       2.0       3.0       4.0        5.0       6.0       7.0 
Time  of  Reverberation— Seconds 


8.0       9.0 


FIG.  8. 


Percentage  Articulation   Curves   for  Rooms   of  Different   Sizes  and  Different  Times   of 
Reverberation 


which  will  obtain  in  rooms  of  different  sizes  and  different  times  of  reverberation,  for  the 
average  speaker,  without  artificial  amplification.  If  an  articulation  of  75  per  cent  is  re- 
garded as  the  minimal  for  satisfactory  hearing  (see  also  article  29),  it  is  apparent  that 
the  average  speaker  will  not  be  heard  satisfactorily  in  an  auditorium  larger  than  about 
1,000,000  cu  ft,  no  matter  what  condition  of  reverberation  has  been  provided  for  the  room. 
Also,  in  an  auditorium  having  a  volume  of  100,000  cu  ft  the  time  of  reverberation  should 
not  exceed  2.70  sec.  The  curves  in  Fig.  8  are  based  upon  the  assumption  that  the  noise 
level  has  been  reduced  to  30  db,  an  unusually  quiet  room,  and  that  k,  =  1.0.  The  need 
for  artificial  amplification  of  speech  in  large  rooms  is  apparent;  amplification  should  be 
provided  in  all  rooms  larger  than  about  50,000  cu  ft,  and  even  in  smaller  rooms  when 
considerable  noise  is  present. 

The  curves  in  Fig.  8  apply  to  the  average  speaker.  Speakers  with  weak  voices  will  not 
be  heard  so  well,  as  is  indicated  by  the  curves,  and  speakers  with  loud  voices  will  be 
heard  better. 

In  all  the  curves  in  Fig.  8  it  will  be  seen  that  the  articulation  is  a  maximum  for  a  particu- 
lar time  of  reverberation,  and  that  this  optimal  time  of  reverberation  increases  with  the 
size^of  the  room,  a  fact  which  is  well  established  by  experience.  Furthermore,  the  percent- 
age'articulation  in  Fig.  8  is  the  average  value;  in  general,  the  articulation  diminishes  pro- 
gressively with  the  increase  of  distance  from  the  speaker. 


32.  MUSIC  ROOMS 

The  reverberatory  properties  of  a  room  are  of  even  greater  significance  for  music  than 
they  are  for  speech.  The  acoustical  properties  of  a  music  room  are  no  less  important  than 
those  of  the  musical  instrument  to  be  played  in  that  room;  indeed,  the  room  and  instru- 


MUSIC  EOOMS 


12-75 


ment  together  comprise  a  coupled  system,  and  it  is  this  combined  system  that  the  ear  or 
microphone  "hears."  The  resonant  frequencies  of  a  room,  considered  in  article  20,  depend 
on  the  dimensions  of  the  room;  their  intensities  and  their  rates  of  growth  and  decay  are 
largely  influenced  by  the  distribution  of  the  absorptive  and  reflective  materials  over  the 
boundaries  of  the  room. 

A  music  room  should  be  so  dimensioned,  shaped,  and  treated  with  absorptive  and 
reflective  materials  as  to  support  and  enhance  the  rich  quality  of  the  individual  tones  and 
harmonies  of  music,  and 
to  join  together  these 
separate  tones  and  har- 
monies so  that  they  coa- 
lesce into  a  continuously 
flowing  melody.  The 
best  music  rooms,  like 
the  best  violins,  are  those 
which  are  free  from 
prominent  resonances 
and  which  have  a  rela- 
tively uniform  steady- 
state  response  through- 
out the  entire  frequency 
range.  Rooms  in  which 
the  ratio  of  height, 
width,  and  length  is 
approximately  2:4:5, 
and  in  which  furred-out 
wood  paneling  and  wood 
flooring  on  wood  joists 
comprise  most  of  the 
interior  boundaries,  are 
found  to  meet  these 
conditions  and  usually  are  highly  acclaimed  by  both  performers  and  listeners.  The  use 
of  non-parallel  pairs  of  opposite  walls,  of  poly  cylindrical  diffusers  of  wood  veneer, 
similar  to  those  illustrated  in  Fig.  3,  and  of  "patches"  of  absorptive  materials 
distributed  over  walls  and  ceiling  so  as  to  give  an  ergodic  diffusion  of  sound  has  given 
good  results. 

The  design  of  music  rooms  should  always  be  guided  by  the  principles  of  wave  acoustics. 
However,  the  optimal  reverberation  characteristics  can  be  best  calculated  by  means  of 

the  approximate  formulas  of  geo- 
metric acoustics,  such  as  eq.  (3) 
of  article  19. 

The  optimal  reverberation 
time  for  music  rooms  depends 
not  only  on  the  size  of  the  room 
but  also  on  the  type  of  music  to 
be  performed  in  the  room.  The 
ideal  arrangement  should  pro- 
vide for  adjustable  reverberation 
so  that  the  optimal  reverberatory 
properties  can  be  readily  ob- 
tained for  all  musical  perform- 
ances for  which  the  room  is  de- 
signed. 

The  chart  in  Fig.  9  shows  the 
optimal  times  of  reverberation 
for  both  speech  rooms  and  music 
rooms.  This  chart  applies  for 


Go.  Ft  12,500     25,000     50,000    100,000  200,000    400,000  800,000 

Volume  of  Room 
FIG.  9.     Optimal  Reverberation  Times  for  Speech  and  Music  Booms 


3.0 
2  0 

Music 
Speech 

\ 

1.0 
0 

^ 

:^ 

*^n 

^^ 

Speech 
Music 

64         128        256        512       1024     2048      4096 
Frequency-Cycles  per  Second 

FIG.  10.  Optimal  Reverberation  Times  at  Different  Fre- 
quencies, for  Speech  and  for  Music,  When  the  Optimal 
Reverberation  Time  at  512  Cycles  Is  1.3  Seconds.^  Similar 
curves  should  be  used  when  the  optimal  time  at  512  cycles 
differs  from  1.3  seconds. 


a  frequency  of  512  cycles.  As  for  speech,  the  reverberation  time  at  128  cycles  should  be 
approximately  25  to  50  per  cent  longer  than  the  time  for  512  cycles;  the  reverberation 
time  should  remain  approximately  constant  for  frequencies  between  512  and  2048  cycles, 
and  should  increase  slightly  for  frequencies  above  204S  cycles  (see  Fig.  10). 

Experience  has  shown  that  the  most  satisfactory  reverberation  time  for  radio 
broadcasting  or  sound-recording  studios  is  about  two-thirds  to  three-fourths  of  the 
accepted  time  for  speech  or  music  rooms  (see  also  Section  16,  Sound-reproduction 
Systems) . 


12-76  ACOUSTICS 

33.  PRACTICAL  PROCEDURE  FOR  OBTAINING  GOOD  ACOUSTICS 

IN  BUILDINGS 

The  procedure  for  obtaining  good  acoustics  in  buildings  begins  with  the  selection  of  the 
site  and  ends  with  the  furnishing,  testing,  and  maintaining  of  the  building.  The  necessary 
steps,  approximately  in  chronological  order,  are  as  follows: 

1.  Selection  of  a  suitable  site  (sound  studios,  theaters,  schools,  churches,  and  hospitals, 
especially,  should  be  located  in  quiet  surroundings). 

2.  Noise  survey,  at  the  proposed  site,  to  determine  the  amount  of  insulation  required  to 
reduce  the  noise  level  in  the  building  to  a  satisfactory  point. 

3.  Insulation  against  outside  noise,  which  includes  not  only  the  selection  of  the  proper 
sound-insulative  and  sound-absorptive  constructions  but  also  the  proper  arrangement  of 
rooms,  corridors,  entrances,  windows,  landscaping,  and  other  appurtenances  of  the  building. 

4.  Design  of  the  shape  of  the  room  (shapes  should  be  designed  which  not  only  will  avoid 
such  acoustical  defects  as  echoes,  interfering  reflections,  room  nutter,  and  sound  foci,  but 
also  will  facilitate  the  most  advantageous  flow  of  diffuse  sound  energy  to  all  auditors  in  the 
room,  and  at  the  same  time  will  preserve  or  even  enhance  the  natural  beauty  of  speech  and 
music). 

5.  Control  of  the  noise  within  the  building,  including  solid-borne  as  well  as  air-borne  noise 
and  vibration. 

6.  Selection  and  distribution  of  the  absorptive  and  reflective  materials  to  provide  the  optimal 
conditions  for  both  steady-state  and  transient  sounds  throughout  the  room,  a  problem  which 
deserves  special  study  and  careful  planning,  and  one  which  involves,  besides  the  acoustical 
characteristics  of  the  materials,  such  properties  as  structural  strength,  decorative  possi- 
bilities, adaptability  to  the  surfaces  available  for,  or  requiring,  absorptive  treatment, 
maintenance,  sanitation,  ease  of  application,  fire  hazard,  absorption  of  water,  attraction 
for  vermin,  "fool-proof ness,"  durability,  and  cost. 

7.  Supervision  of  the  installation  of  acoustical  materials  (especially  necessary  for  the 
application  of  acoustical  plaster — in  large  buildings  it  is  advisable  to  require  the  plastering 
contractor  to  prepare  a  small  room  for  test  and  approval  before  the  plaster  is  used  in 
other  parts  of  the  building). 

8.  Installation  of  high-quality  amplifying  equipment  under  the  supervision  of  a  competent 
engineer  is  necessary  in  all  large  auditoriums;  even  in  rooms  seating  as  few  as  200  or  300 
persons  it  will  be  found  that  many  speakers  have  weak  voices  that  require  amplification. 

9.  Inspection  of  the  finished  building  should  include  tests  to  determine  whether  the 
sound  insulation,  the  sound  absorption,  and  the  other  acoustical  properties  have  been 
satisfactorily  attained. 

10.  Maintenance  instructions,  preferably  in  loriting,  should  be  left  with  the  building  man- 
ager, indicating  (a)  how  the  acoustical  materials  can  or  cannot  be  cleaned  or  redecorated, 
(6)  which  furnishings  in  the  building  are  essential  to  good  acoustics,  and  (c)  how  the 
humidity  of  large  speech  and  music  rooms  should  be  maintained  in  order  to  avoid  excessive 
absorption  of  high-pitched  sounds. 

The  foregoing  steps,  or  their  equivalent,  if  carefully  executed,  will  lead  to  good  acoustics. 
Developments  in  modern  theories  of  room  acoustics,  supplemented  by  additional  empirical 
data,  will  contribute  to  more  reliable  criteria  than  are  now  available  for  determining  the 
best  acoustical  shape  of  a  room  and  the  most  favorable  distribution  of  absorptive  and  re- 
flective materials  throughout  the  room;  but  if  proper  use  is  made  of  what  is  now  known 
there  need  be  no  anxiety  respecting  the  outcome  in  the  acoustics  of  buildings — the  out- 
come will  be  good. 

BIBLIOGRAPHY 

Bagenal  and  Wood,  Planning  for  Good  Acoustics.     Methuen/ London  (1931). 

Davis  and  Kaye,  The  Acoustics  of  Buildings.     G.  Bell  and  Sons,  London  (1927). 

Fleming  and  Allen,  Modern  Theory  and  Practice  in  Building  Design,  The  Institution,  London  (1945). 

Fitzmaurice  and  Allen,  Sound  Transmission  in  Buildings,  His  Majesty's  Stationery  Office,  London 

(1939) . 
Glover,  Practical  Acoustics  for  the  Constructor.    Chapman  and  Hall,  London  (1933).     (Contains  a 

complete  bibliography  of  books  and  journal  articles.) 
Knudsen,  V.  O.     Architectural  Acoustics.     John  Wiley  (1932), 
Knudsen  and  Harris,  Acoustical  Designing  in  Architecture.    John  Wiley  (1949). 
Rettinger,  Applied  Architectural  Acoustics,  Chemical  Publishing  Co.  (1947). 
Sabine,  P.  E.,  Acoustics  and  Architecture.     McGraw-Hill  (1932). 
Strutt,  Raumakustik.     Handbuch  der  Bxperimentalphysik,  Bank  XVII/2.     Akad.  Verlagsgesellschaft, 

Leipzig  (1933).  *^  ' 

Watson, F.  IL,  Acoustics  of  Building*.     John  Wiley  (193S). 


SECTION  13 
ELECTROMECHANICAL-ACOUSTIC  DEVICES 


EFFECTS  OF  THE  ACOUSTIC  MEDIUM 
BY  HUGH  S.  KNOWLES 

ART.  PAGE 

1.  Physical  Properties  of  Common  Acoustic 

Media 02 

2.  Mechanical    Impedance    to    Motion   of 

Some  Simple  Acoustic  Radiators 03 

3.  Horns 05 


LOUDSPEAKERS 
AND  TELEPHONE  RECEIVERS 

BY  HUGH  S.  KNOWLES 

4.  Acoustic  Radiators 08 

5.  Efficiency 10 

6.  Moving-conductor  Speakers 11 

7.  Magnetic-armature  Speakers 15 

8.  Condenser  Speakers 16 

9.  Pneumatic  Speakers 17 

10.  Telephone  Receivers  (Earphones) 17 

11.  Performance  and  Tests 18 


MICROPHONES 

BY  HUGH  S.  KNOWLES 

12.  Force  on  the  Microphone 22 

13.  Moving-conductor  Microphones 23 

14.  Condenser  Microphones 24 

15.  Magnetic-armature  Microphones 25 

16.  Crystal  Microphones 25 

17.  Carbon  Microphones 26 

18.  Directional  Characteristics 26 

19.  Performance  and  Tests 26 


MAGNETIC  RECORDING 

AND  REPRODUCING  OF  SOUND 

AnT     BY  L.  VIETH  and  H.  A.  HBNNING     PAGB 

20.  Erasing,    Recording,    and    Reproducing 

Arrangements 28 

21.  Erasing,    Recording,    and    Reproducing 

Processes 29 

22.  Recording  Media 35 

MECHANICAL  RECORDING 

A1*D  REPRODUCING  OF  SOUND 

BY  L.  VIETH  and  H.  A.  HENXING 

23.  Recording  Instruments 37 

24.  Recording  and  Reproducing  Media, 40 

25.  Reproducing  Instruments 43 

26.  Sources  of  Distortion 45 

PHOTOGRAPHIC  SOUND  RECORDING 

BY  C.  R.  KEITH 

27.  Light-valve  Recording  System 48 

28.  Refiecting-galvanometer  Recording  Sys- 

tem      50 

29.  Flashing-lamp  and  Kerr  Cell  Recording 

Systems 51 

30.  Sound-on-film  Reproducing  Systems 52 

PIEZOELECTRIC  CRYSTALS 
BY  W.  P.  MASON 

31.  Definition  of  Effects 55 

32.  Application  of  Piezoelectric  Crystals  ...     58 

33.  Properties  of  Quartz 58 

34.  Properties  of  RocheEe  Salt 65 

35.  Properties    of   Ammonium    Dihydrogen 

Phosphate  (ADP) 68 


13-01 


ELECTROMECHANICAL-ACOUSTIC  DEVICES 
EFFECTS  OF  THE  ACOUSTIC  MEDIUM 

By  Hugh  S.  Knowles 

"An  electroacoustic  transducer  is  a  transducer  which  is  actuated  by  power  from  an 
electrical  system  and  supplies  power  to  an  acoustic  system,  or  vice  versa."  (I.R.E. 
Standards.) 

The  theory  of  operation  of  electroacoustic  transducers  is  an  extension  of  the  theory  of 
electromechanical  transducers,  in  which  account  is  taken  of  the  reaction  of  the  fluid,  or 
acoustic  medium,  on  the  diaphragm.  (See  Section  5,  article  33.) 

1.  PHYSICAL  PROPERTIES  OF  COMMON  ACOUSTIC  MEDIA 

The  velocity  of  a  sound  of  small  (infinitesimal)  wave  amplitude  depends  on  the  elas- 
ticity and  density  of  the  fluid.  The  velocity  of  propagation,  c,  of  a  sound  wave  is 


cm  sec-*  (1) 

where  k  is  the  volume  modulus  of  elasticity,  p  is  the  density  in  grams  cm~3,  7  is  the  ratio 
of  the  specific  heat  at  constant  pressure  to  that  at  constant  volume,  and  PQ  is  the  static 
pressure  of  the  fluid. 

The  characteristic  impedance,  ZQ  (also  sometimes  called  surge  impedance  or  acoustic 
resistance)  ,  of  the  medium  is 

ZQ  =*  V&p  =  pc     mechanical  ohms  em~2  (2) 

1.  Air.     The  density,  p,  at  20  deg  cent  and  po  =  760  mm  (  =  10s  dynes  cm"2)  is  0.001205 
gram  cm""3;  y  ==  1.41,  giving 

c  =  33,060  -h  610     cm  sec"1  (3) 

where  6  is  the  temperature  in  degrees  centigrade.    Also  the  radiation  resistance  of  air  is 

ZQ  =  42.8  —  0.0790  —  41.2     mechanical  ohms  cm^2  at  20  deg  C  (4) 

The  pressure  level,  Lp,  in  decibels  is 

LP  -  20  logic  -  =  74  +  20  logioj>  (5) 

Po 

where  p  is  the  pressure  and  pp  the  reference  pressure  of  0.0002  dyne  cm"2.  The  intensity,  I, 
in  the  direction  of  propagation,  of  a  plane  or  spherical  "free"  (i.e.,  no  reflections)  sound 
wave  is 

_2 

/  =  ^-  =  2.42  X  10  "V    watt  cm-2  (6) 

PC 

where  p  is  the  rms  sound  pressure  in  dynes  cm"2. 

2.  Hydrogen.     The  density,  p,  of  hydrogen  at  0  deg  cent  and  at  po  =  760  mm  is  approxi- 
mately 0.00009  gram  cm~3.    The  velocity  c  =  1.26  X  106  cm  sec"1,  from  which  ZQ  =  11 
mechanical  ohms  cm~2. 

3.  Water.     The  density,  p,  of  water  is  =  1.0  gram  cm""3.    The  value  of  c  at  20  deg  cent 
=  1.46  X  105  cm  sec"1,  from  which  ZQ  —  1.46  X  105  mechanical  ohms  cm"2. 

REACTION  OF  ACOUSTIC  MEDIUM  ON  A  DIAPHRAGM.  The  audible  frequency 
range  of  sounds  ©overs  roughly  10  octaves.  Even  the  more  important  range  from  80  to 
8000  cycles  covers  nearly  7  octaves,  giving  a  wavelength  range  of  roughly  428  cm  (14.1  ft) 
to  4.28  cm  (1.69  in.).  Radiation,  diffraction,  and  reflection  phenomena  which  depend  on 
the  relative  length  of  the  sound  wave  and  the  linear  dimensions  of  the  radiator,  or  collector, 
therefore  differ  greatly  in  different  portions  of  the  wavelength  range,  and  hence  simplifying 
assumptions  can  ordinarily  be  made  only  over  restricted  frequency  ranges. 

13-02 


MECHANICAL  IMPEDANCE  TO  MOTION 


13-03 


2.  MECHANICAL  IMPEDANCE  TO  MOTION  OF  SOME 
SIMPLE  ACOUSTIC  RADIATORS 


The  impedance  to  motion  of  a  diaphragm  is  altered  by  its  contact  with  the  acoustic 


. 

SOURCE  RADIATING  PLANE  SOUND  WAVES.  The  impedance  per  unit  area  in 
contact  with  the  fluid  is  the  characteristic  impedance  of  the  medium,  and  the  total  increase 
in  mechanical  impedance  to  motion  of  a  radiator  of  effective  area,  S,  resulting  from  contact 
with  the  fluid  medium,  is  » 

*/  -  rf  =  *bS  (7) 

or  =  41.2  X  S  mechanical  ohms  hi  the  case  of  ah*.   At  any  frequency,  the  radiated  acoustic 
power  is 

Pt  =  3VO-'  (S) 

or  4.12  X  10  -6s2  X  watts  in  the  case  of  air  (s  is  the  velocity  of  the  radiator  element). 

PULSATING  SPHERE  RADIATING  SPHERICAL  SOUND  WAVES  INTO  AN 
UNLIMITED  ACOUSTIC  MEDIUM.  A  pulsating  sphere  is  one  in  which  the  surface 
vibrates  or  pulsates  with  small  amplitude  and  uniform  velocity  in  a  radial  direction.  At 
any  frequency  the  impedance  per  unit  area  in  mechanical  ohms  is 


ZA'  =  TA'  +  j% 


pc 


pR 


(9) 


where  k  =  2?r/X  =  w/c  =  2-n-f/c,  where  X  is  the  length  of  the  emitted  sound  wave,  and  R 
is  the  radius  of  the  sphere  in  centimeters.  This  leads  to  an  equivalent  circuit  of  parallel 
mass  and  resistance  elements  of  values  (for  the  whole  surface)  m/  —  4xpfi3  grams  and 
Tf  =  4.7rR2pc  mechanical  ohms. 

Graphical  plots  of  TA'  and  XA'  are  given  in  Fig.  1.    The  imaginary  or  reactance  term  is 
in  phase  with  the  acceleration  and  is  an  inertia  or  mass  reactance  term. 


so 

1 

~\ 

50 
40 
30 

|§2° 
'o  >-     8 

^ 

s~+* 

S 

[7^ 

' 

/ 

f 

\ 

\ 

N 

mpedance  per  Sq  Cm 
per  Sq  Cm  per  Cm  p 

co  M  to  w  £kuicn 

-jAt- 

-* 

i 

/  \ 

'i 

•^ 

*j 

r 

) 

\J 

k 

^^ 

/ 

/ 

f 

1 

x> 

\f\^ 

V 

y 

^ 

'/ 

\ 

\ 

0)    03     .6 

=f  i 
Is  i 

.2 
.0 

^jX- 

\ 

// 

1 

/ 

y 

^ 

.' 

1 

^ 

^ 

/ 

1  1 

/ 

Dl   .002     ,004         .01      .02     .04     .08.1       .2    ,3.4   .6.81.0     2     3   4     6  S  10      2030       60    10 
KR=1^=«S. 

FIG.  1.     Air  Resistance  and  Reactance  per  Unit  Area  on  One  Side  of  a  Pulsating  Sphere  of  Radius 
R(T'A  and  X'A)  aad  on  One  Side  of  a  Circular  Piston  of  Radius  R  (r  A  and  XA)  in  an  Infinite  Baffle 

The  fluid  increases  the  impedance  to  motion  of  the  spherical  surface  or  diaphragm  by  an 
amount  z/  =  SZA'.  At  any  frequency  the  radiated  acoustic  power  is  Pf  =  IV/  X  10 ~7  watt. 

Special  Case  of  a  "Point**  Source  (R  =  0).  When  the  length  of  the  sound  wave  is 
large  in  comparison  with  the  radius  of  the  sphere  (\^>  R  and 


=  lirpR*  — 


mechanical  ohms 


(10) 


13-04  ELECTKOMECHANICAL-ACOUSTIC  DEVICES 

CIRCULAR  RIGID  DIAPHRAGM  OR  "PISTON"  reciprocating  or  vibrating  sinus- 
oidally  in  a  (perfect-fitting)  cylindrical  hole,  in  an  infinite  P^?7^f  °f^™%^ra^*^5 
into  an  unlimited  fluid  ('  'semi-infinite"  medium)  on  each  side  of  the  baffle,  is  another 
configuration  of  interest.  (R  is  the  radius  in  centimeters.) 

The  impedance  per  unit  area  is  not  constant  over  the  surface  in  this  case,  but  since  the 
radiator  is  assumed  rigid  an  average  value  may  be  taken.  At  any  frequency  the  impedance 
in  mechanical  ohms  cm"2  is  given  by  ZA  =  rA  +  jxA  as  plotted  in  .big.  1. 

Since  the  imaginary  term  is  in  phase  with  the  acceleration,  it  is  a  mass  reactance  term. 
The  increase  in  mechanical  impedance  to  motion  of  a  piston,  z/  =  rf  +  yxf  -  £>(fA  -rJXA), 
which  results  from  an  air  "load"  may  be  obtained  directly  fromthese  curves  by  multiply- 
ing the  ordinate  for  any  value  of  kR  by  the  area  of  the  piston  When  the  fluid  is  air,  piston 
radiators  are  usually  operated  with  fluid  on  both  sides  of  the  baffle.  Both  sides  of  the 
piston  must  then  be  considered,  in  which  case  S  -  2ir#.  The  total  apparent  increase  in 
mass  of  the  piston  is  mf  =  SXA/M-  At  any  frequency,  the  radiated  acoustic  power  is 

P/Special  Case  ofT^Point"  Source  (R  -  0).    When  the  length  of  the  sound  wave  is 
large  in  comparison  with  the  radius  of  the  piston,  the  impedance  per  unit  area  is 

.  ju pR  (11) 


Frequency,  CPS  for  12  Speaker  (10^  Piston) 


5O       60  70  80     100 


300        400 


\ 


\\ 


Reference  to  Pig.  1  will  show  that,  for  small  values  of  kR,  xA^>rA,  so  that  ZA  =  JXA. 
That  is   the  increase  in  impedance  to  motion  of  the  piston,  which  results  from  contact 
with  the  fluid,  is  largely  reactive.    At  this  low-frequency  condition  the  radiation  imped- 
ance of  the  piston  may  be  represented  by  an  equivalent  circuit  of  parallel  mass  and  resist- 
ance elements  of  values  8pfi»/3  grams  and  (128/9^)  wR*pc  mechanical  ohms  respectively. 
The  values  are  O.OOCSed3  grams  and  SOld2  mechanical  ohms,  d  being  the  piston  diameter 
in  niches.  .     .    ... 

It  has  been  found  experimentally  that  when  a  conventional  rigid  conical  diaphragm 
loudspeaker  is  used  S  is  approximately  the  area  of  the  "base,"  or  large  end,  of  the  cone; 

also,  that  the  usual  magnetic 
structure  changes  the  radiation 
from  the  rear  surface  of  the 
cone  by  a  negligible  amount  at 
low  frequencies. 

MULTIPLE  PISTON'S. 
When  more  than  one  piston 
radiates  into  a  common  region 
in  the  medium  any  one  position 
experiences  not  only  the  force 
on  its  surface  arising  from  its 
own  vibration  but  additional 
forces  due  to  the  vibration  of 
the  other  pistons.  The  result- 
ant force  is  the  vector  sum  of 
the  individual  forces.  The  mag- 
nitude and  phase  of  each  force 
depend  on  the  diameter,  veloc- 
ity, and  frequency  of  vibration 
of  the  piston  giving  rise  to  the 
force  and  upon  its  distance  from 
the  reference  piston.  The  ratio 
of  the  resultant  force  on  any 
piston,  to  its  velocity,  is  the 
PIG.  2.  Ratio  of  Total  Radiation  Resistance  of  a  Piston,  Vibrat-  total  fluid  or  radiation  imped- 
KpS^e^elSe%^nR^^effSATfto  ««*  ""A  comprises  the  self- 
tangent  pistons.  Curve  3,  each  of  four  tangent  pistons  in  square  impedance  due  to  its  own  vi- 
array.  Curve  4,  each  outer,  and  Curve  5,  each  inner,  piston  of 
four  tangent  pistons  in  a  straight  line.  Curve  6,  each  of  two  pis- 
tons with  centers  three  diameters  apart.  All  pistons  have  equal 
velocities  (both  magnitude  and  phase)  and  equal  diameters,  and 
vibrate  in  an  infinite  plane  baffle. 


=  2 


01 

0.1 


1 


\ 


0.15       0.2 


0.3      o.4 
2TR  _  CJR 
X  c 


0.5  0.60.70.8    i.o 


bration,  discussed  above,  and 
the  mutual  impedances  due  to 
the  vibration  of  the  other  pis- 
tons. From  Fig.  2  it  will  be 


seen  that  the  real  part  of  the 

total  impedance  increases  as  the  distance  between  the  pistons  decreases  and  their  num- 
ber and  hence  total  area  increase.  When  the  pistons  are  close  together,  vibrate  with 
equal  amplitude,  and  are  separated  by  a  small  fraction  of  a  wavelength,  they  approximate 
a  single  piston  equal  in  area  to  the  combined  area  of  the  individual  pistons.  It  may  also 


HORNS 


13-05 


be  seen  that  as  the  separation  and  frequency  are  increased  the  phase  of  the  force  arising 
from  the  vibration  of  the  other  piston  is  retarded  and  may  give  rise  to  a  component  out  of 
phase  with  the  velocity  of  the  reference  piston  which  lowers  the  radiation  or  fluid  resistance 
below  the  value  the  reference  piston  would  have  alone.  This  occurs  in  curve  6,  Fig.  2, 
for  values  of  &R/c  greater  than  0.5. 

ENCLOSED  BACK  PISTON.  Same  as  preceding  case  but  with  small  enclosure  to 
suppress  radiation  from  one  side  of  a  diaphragm,  or  to  add  stiffness  to  the  vibrating 
system. 

The  fluid  impedance  at  the  external  surface  corresponds  to  the  preceding  case  (note 
that  S  =  -n-R*). 

If  the  dimensions  of  the  enclosure  are  larger  than  the  diameter  of  the  piston  the  en- 
closure adds  approximately  the  same  effective  mass  per  unit  area,  mj,  as  the  unlimited 
fluid  medium.  Therefore,  the  effective  area  used  in  calculating  the  total  effective  fluid 
mass  is  S  =  2?r.R2,  that  is,  m/  =  2TrJ&m& . 

If  the  length  of  the  radiated  sound  wave  is  roughly  four  or  more  times  the  maximum 
linear  dimension  of  the  enclosure,  uniform  adiabatic  compression  of  the  fluid  occurs.  The 
enclosure  then  increases  the  stiffness  (see  Section  5,  article  33)  per  unit  area  of  the  mechan- 
ical system  by  an  amount 


=  -=-    cm  dyne"1  cm"2 

y  o 


(12) 


(Vo  is  the  volume  of  the  enclosure).  The  total  increase  in  stiffness,  s/,  due  to  the  fluid, 
is  SSA. 

The  enclosure  does  not  approximate  a  constant  stiffness  when  the  length  of  the  sound 
wave  is  less  than  four  times  the  maximum  linear  dimension  of  the  enclosure. 

If  the  purpose  of  the  enclosure  is  to  provide  a  "sink"  to  absorb  back  side  radiation,  ab- 
sorbing material  is  usually  placed  hi  the  enclosure.  This  increases  the  effective  resistance 
of  the  vibrating  system.  The  absorption  coefficient  of  the  material  used  is  normally  high 
enough  at  the  high  resonant  frequencies  of  the  enclosure  to  make  the  enclosure  approxi- 
mate a  semi-infinite  medium. 

3.  HORNS 

"A  horn  is  an  acoustic  transducer  consisting  of  a  tube  of  varying  sectional  area." 

The  proper  use  of  a  horn,  as  the  radiating  portion  of  a  loudspeaker  leads  to  better  control 
of  the  response,  efficiency,  and  directional  characteristics.  In  addition  these  character- 
istics may  be  controlled  almost  independently  of  one  another. 

The  total  radiation  response  of  a  horn  considered  as  an  acoustical  circuit  element  is 
determined  largely  by  its  throat  impedance  as  a  function  of  frequency.  In  a!  well- 
designed  horns,  transmission  losses  are 

ininimized,  so  that  the  energy  output  is  Diameter  d 

closely  equal  to  the  input. 

The  rigorous  calculation  of  the  throat 
impedance  is  possible  for  but  very  few 
useful  horn  contours,  and  so  approxi- 
mate methods  are  used.  The  low-fre- 
quency region  is  of  greatest  interest  to 
the  horn  designer,  as  all  horns  have  a 
high-frequency  throat  mechanical  im- 
pedance which  approaches  a  constant 
resistance  that  is  the  same  (per  unit 
area)  for  all  horns.  Experiment  reveals 
that  the  low-frequency  wave  fronts  in 
the  horn  are  smoothly  curved  surfaces 
(Fig.  3) ;  by  expressing  this  mathemati- 
cally there  results  a  pressure  wave  equa- 


FIG.  3.    Low-frequency  Wave  Fronts  in  Long  Straight 
Axis  Horn  of  Circular  Croes-sectim 


tion  in  which  the  only  space  variable  is  the  axial  distance.  For  convenience,  all  horn- 
design  work  is  usually  referred  to  a  straight  axis  horn  of  circular  cross-section,  as  in  Fig.  3. 
The  question  arises  as  to  what  types  of  horns  have  this  kind  of  simplified  behavior. 
Starting  from  the  experimental  data  it  is  observed  that  the  "chords"  (the  diameter,  d, 
in  Fig.  3)  for  the  wave  fronts  are  approximately  proportional  to  the  square  root  of  the 
area  of  the  wave  front,  provided  that  the  expansion  of  the  horn  is  not  too  rapid.  If  the 
sound  pressure  is  assumed  to  decrease  steadily  as  the  horn  expands,  modified  by  the 
change  of  phase  down  the  horn,  this  assumption  may  be  expressed  analytically  and  in- 
serted into  the  pressure  wave  equation  together  with  the  relation  between  d  and  the  area 


13-06  ELECTROMECHANICAL-ACOUSTIC  DEVICES 


7=0 


of  wave  front.    This  leads  to  a  relation  between  d  and  x,  the  axial  distance,  which  when 
solved  yields 

d  =  dt[cosh  (X/XQ)  +  Tsinh  (x/xo)]  (13) 

Here  dt  is  the  diameter  at  the  throat;  x0  is  a  reference  distance  fixing  the  rate  of  taper  of 

the  horn,  and  is  related  to  the  cutoff  fre- 
quency fe  by  fe  =  c/2ira:o;  and  T  is  a  param- 
eter by  which  a  particular]  horn  contour  is 
selected.  The  names  "catenoidal  horns" 
and  "Salmon  horns"  have  been  suggested 
for  this  family,  the  latter  after  the  person 
who  first  described  their  characteristics. 

The  family  is  the  most  general  one  con- 
sistent with  the  simplifying  assumptions 
made;  f  or  T  =  1  there  results  the  familiar 
exponential  horn  d  =  dt  exp  (re/rco),  while 
for   T  -  oo  there   results,   by   a  limiting 
_    process,    the    conical    horn.     At    T  =  0, 
,0,    d  =  dt  cosh  (X/XQ),  which  may  be  termed 
the  cosh  horn.    Since  this  family  includes 
the  most  widely  used  horn  contours,  it  will 
be  taken  as  a  basis  of  discussion.     Contours  for  T  =  0,  1,  5,  and  °o  are  shown  in  Fig.  4. 
The  low-frequency  throat  impedance  of  all  practical  horns  shows  considerable  variation 
with  frequency  due  to  reflections  from  the  mouth.     Hence  it  is  common  engineering 

practice  to  state  as  the  throat  imped- 

2.0  ,  —  ,  _  ,  -  —  i  -  1      ance  that  for  the  infinite  horn  (outgoing 

wave  only);  in  practice  the  actual  im- 
pedance varies  about  the  infinite  horn 
value  as  a  mean,  or  trend,  which  is  ap- 
proached as  reflections  from  the  mouth 
are  decreased.  When  the  mechanical 
throat  impedance  is  evaluated  for  the 
horns  of  eq.  (13)  there  results 

Zt  =  Tt 


:  Stpc 


FIG.  4.    Contours  of  Horns  of  Eq.  (13)  for  T  •• 
1,  5,  and  to 


([i-  Qfc/OT 
1     i  -  d  - 


, . 

(a)  Throat  Resistance 


0.2 


(14) 

where  St  is  the  throat  area;  pc  is  the 
characteristic  impedance  of  the  me- 
dium; the  cutoff  frequency  /«  =  C/ZTTXQ, 
c  being  the  velocity  of  a  free  sound 
wave.  An  examination  of  eq.  (14)  re- 
veals that  rt  is  zero  for  f  <  fe  (for  the 
infinite  horn)  ,  so  that  below  fc  the  im- 
pedance is  entirely  reactive.  Figure  5 
shows  rt/Stpc  and  xt/Stpc  as  functions 
of  f/fc  for  T  =  0,  0.5,  1,  and  5.  The 
equivalent  circuit  of  the  mechanical  im- 
pedance of  eq.  (14)  is  as  shown  in 
Fig.  6;  note  that  the  parallel  elements 
are  simple;  only  m*  varies  with  Z7,  and 
only  rt  varies  with  /. 


f/fc 


10 


Throat  Reactance 


FIG.  5.     Throat  Impedance  of  Infinite  Horns  of  Eq. 
(13).    The  fc  in  the  abscissa  is  the  cut-off  frequency. 


FIG.   6.     Equivalent   Circuit   of   Throat   of 

Horns  of  Eq.  (13) .    Negligible  reflection  from 

mouth. 


The  selection  of  the  optimum  member  of  the  above  family  of  horns  cannot  be  considered 
apart  from  the  loudspeaker  unit  or  other  source  out  of  which  the  horn  is  to  work,  as  both 


HORNS 


13-07 


form  a  fairly  tightly  coupled  system.     The  essential  mechanical  elements  are  shown  in 
Fig.  7  for  a  moving-coil  driving  unit  as  discussed  in  article  6. 

The  electrical  system  consists  of  a  generator  of  constant  emf ,  e,  and  total  electrical 
impedance,  r«,  which  is  here  assumed  resistive  and  equal  to  the  generator  resistance  proper 
plus  the  blocked  resistance  of  the  voice  coil.  The  resulting  force  input  to  the  circuit  of 
Fig.  7  is  thus  $le/ret  which  is  fairly  constant  for  horn  loudspeakers.  As  seen  from  the  me- 
chanical mesh,  the  electrical  side  also  contributes  a  "source  impedance"  n  =  (£kVr.  which 
is  as  constant  as  the  force.  In 
these  expressions  /S  is  the  mag-  /22z2 

netic  flux  density  in  the  gap 
and  I  is  the  conductor  length. 

In  the  purely  mechanical  sys- 
tem mi  is  the  mass  of  the  loud- 
speaker motor,  and  si  its  stiff- 
ness, which  may  include  air 
trapped  back  of  the  diaphragm. 
The  stiffness  of  the  air  sz,  of 
volume  V,  between  diaphragm 
and  horn  throat  (the  sound 
chamber),  is  equal  to  pc?S<?/V 


I — VW O^MJLy — {(• 

O  *i  7ft  S 


Force=s 


f 


FIG.  7.    AI. 
speaker.     (, 


te  Equivalent  Circuit  of  Typical  Horn  Loud- 
is  the  impedance  ratio  of  the  fluid  trans- 
former formed  by  the  areas  84  and  St. 


when  referred  to  the  diaphragm  area  Sd-  Because  Sd  is  usually  different  from  Si  the  horn 
throat  elements  of  Fig.  6  appear  at  the  diaphragm  multiplied  by  the  fluid  transformer 
impedance  ratio  (Sd/St)z,  which  is  controlled  by  the  horn. 

The  configuration  of  the  circuit  is  that  of  a  band-pass  filter;  however,  this  is  not  a  satis- 
factory basis  of  design,  as  only  a  half  section  is  present.  In  practice  a  fiat  response  is  not 
always  the  desired  goal,  and  so  the  elements  are  chosen  with  a  particular  application  in 

mind.     Usually  the  unloaded  resonant  frequency,  f\  —  —  (  —  J    ,  of  the  motor  is  placed 

2v  \mi/ 

below  the  midband,  while  the  other  resonant  frequency,  fz  —  —  (  —  j    ,  is  placed  above. 

The  bandwidth  is  fixed  largely  by  ss/Sj;  thus  the  sound-chamber  volume  and  hence  the 
clearance  to  the  diaphragm  should  be  as  small  as  possible  for  a  wide  band.  The  horn 
influences  this  by  controlling  m%  through  the  parameter  T,  permitting  /j  to  be  properly 
placed,  with  a  suitable  $2. 

The  sound  chamber,  which  may  be  regarded  as  a  part  of  the  horn  just  as  a  transformer 
is  often  associated  with  a  loudspeaker,  serves  to  reduce  the  effect  of  mi  at  high  frequencies 

by  the  mi  —  Sa  resonance.  Simi- 
larly at  low  frequencies  the  m±—  si 
resonance,  as  influenced  by  the 
value  of  T,  may  produce  a  rise  in 
response.  This  reactance  annul- 
ling permits  good  low-frequency 
response  to  be  obtained  by  plac- 
ing the  horn  cutoff  frequency,  /c, 
below  /i,  the  resonant  frequency 
of  the  Wi  —  si  combination. 

The  termination  r2  is  usually 
close  to  n  to  obtain  approximate 
matching  in  the  midHresponse 
region,  thus  governing  the  maxi- 
mum efficiency.  This  involves 
the  proper  relation  between  horn- 
throat  area  and  useful  magnetic 
energy  in  the  air  gap.  When 
n  ==  r>,  variations  in  horn-throat 
FIG.  8.  Approximate  Relation  between  Effective  Mouth  Di-  Smrx^nnnp.  Art*  to  month  rofieo- 
ameter  and  Frequency  for  Exponential  Horn  impedance  due  to  moutft  rejec- 

tions have  a  minimum  enect  on 

the  radiated  power.  These  reflections  are  minimised  also  when  the  product  of  mouth 
diameter  in  inches  and  cutoff  frequency  in  cps  is  greater  than  4000;  this  product  may  be 
made  as  little  as  2000  if  r2  =  n- 

Since  diaphragms  are  often  called  on  to  radiate  energy  at  wavelengths  for  which  destruc- 
tive interference  may  take  place  across  the  diaphragm,  it  is  usual  to  remove  the  radiation 
by  one  or  more  annular  slots  so  placed  that  phase  effects  are  minimized.  The  annular 
passages  may  then  be  constricted  in  average  diameter  until  the  circular  horn  section  is 
reached. 


0.1 


10* 


2  x  104 


fd, 


5xl04  105 

,  in  CPS  x  Inches 


2xl05 


13-08  ELECTROMECHANICAL-ACOUSTIC  DEVICES 

The  directional  properties  of  a  horn  are  largely  determined  by  the  ^  mouth  geometry, 
particularly  the  diameter  and  slope.  As  frequency  is  increased,  there  fe  reached  a  value 
at  which  the  emergent  sound  is  so  directional  that  the  wave  does  not  "touch"  the  mouth 
portion  at  all.  Thus  the  effective  mouth  diameter  decreases  as  the  frequency  is  raised, 
and  in  some  designs  the  polar  response  pattern  becomes  almost  independent  of  frequency. 
Data  reported  in  the  literature  may  be  presented  in  the  form  of  equivalent  mouth  diameter 
as  a  function  of  frequency.  In  Fig.  8  this  is  shown  for  one  group  of  measurements  on 
exponential  horns;  the  curves  are  rough  approximations  and  should  be  taken  as  indicating 
only  the  trends.  The  ratio  (dem/dm)  is  that  of  equivalent  and  actual  mouth  diameters. 

In  general,  the  steeper  the  mouth  slope,  the  less  directional  the  horn;  but,  if  this,  generali- 
zation is  carried  too  far,  the  violence  done  to  the  wave  fronts  appears  as  a  rough  response 
characteristic.  The  directional  effect  of  horns  is  not  too  serious  hi  practical  applications, 
for  often  this  property  is  useful  in  reducing  acoustic  feedback  and  improving  the  sound 
level  in  a  desired  localized  region.  When  uniform  radiation  over  a  large  solid  angle  is  re- 
quired, multicell  horns  are  commonly  used.  Often  the  shape  of  the  horn  may  be  altered 
so  as  to  provide  improved  distribution,  and  this  frequently  results  in  a  more  uniform  space- 
response  variation. 

BIBLIOGRAPHY 

American  Standard  Z24.1-1942.     (Terminology.) 

BaUantine,  Stuart,  J.  Franklin  Inst.,  VoL  203,  86  (1927).    (Bessel  horns.) 

Crandall,  I.  B.,  Theory  of  Vibrating  Systems  and  Sound,  p.  85  ff.    (Impedance  of  medium  and  of  horns.) 

Hall,  W.  MM  J.A.S.A.,  Vol.  3,  552  (1932).    (Sound  field  in  horns.) 

Hanna,  C.  R.t  and  J.  Slepian,  Trans.  AJ.E.E.t  Vol.  43,  393  (1924).    (Horn  loudspeakers.) 

Hoersch,  V.  A.t  Phys.  Rev.t  VoL  25,  218  and  225  (1925).    (Transverse  vibrations  in  horns.) 

International  Critical  Tables,  Vol.  4,  453  (1929).    (Constants.) 

I.R.E.  Standards  on  Electroacoustics  (1938).     (Terminology.) 

Klapman,  S.  J.,  J.A.S.A.,  Vol.  11,  289  (1940).    (Multiple  pistons.) 

Lamb,  BL,  Proc.  Roy.  Soc.  London,  VoL  98,  205  (1920).    (Fluid  impedance.) 

McLachlan,  N.  W.,  Loudspeakers  (1934).    (Fluid  impedance,  general,  and  bibliography.) 

McLachlan,  N.  W.,  and  S.  Goldstein,  J.A.S.A.,  VoL  6,  275  (1935).    (Finite  pressure  distortion  horns.) 

Mason,  W.  P.,  B.S.T.J.,  VoL  6t  258  (1927).    (Horns  and  other  acoustic  elements.) 

Phelps,  W.  D.,  J.A.S.A.,  VoL  12,  68  (1940).    (Finite  horn  boundary  impedance.) 

Eayleigh,  J.  W.  S.»  Theory  of  Sound,  VoL  II,  164.    (Fluid  impedance.) 

Salmon,  V.,  J.A.S.A.,  VoL  17,  199  and  212  (1946).    (Horns.) 

Stewart,  G.  W.f  and  Lindsay,  R.  B.,  Acoustics,  p.  132.    (Horns  and  general.) 

Thuras,  A.  L.,  R.  T.  Jenkins,  and  H.  T.  O'Neil,  B.S.T.J.,  VoL  14,  159  (1935).     (Finite  pressure 

distortion  horns.) 

Van  Urk,  T.,  and  R.  Venneulen,  Philips  Tech.  Rev.,  VoL  4,  213  (1939).     (Radiation  admittance.) 
Webster,  A.  G.,  Proc.  Nail.  Acad.  Sci.,  VoL  5,  275  (1919).    (Horns.) 


LOUDSPEAKERS  AND  TELEPHONE  RECEIVERS 

By  Hugh  S.  Knowles 

A  loudspeaker  is  an  electroacoustic  transducer  actuated  by  energy  from  an  electrical 
system  and  radiating  energy  into  an  acoustical  system,  the  spectral  composition  of  the 
energy  in  the  two  systems  being  substantially  equivalent.  Loudspeakers  may  be  classified 
as  to  type  of  radiator  or  radiating  system,  type  of  motor,  and  reversibility. 

4.  ACOUSTIC  RADIATORS 

DIAPHRAGMS.  The  transformation  of  electrical  into  acoustical  energy  is  usually 
accomplished  by  electrically  actuating  a  surface  or  diaphragm  in  contact  with  air,  or  some 
other  fluid,  causing  it  to  move  and  set  the  adjacent  ah*  particles  in  motion.  (See,  however, 
article  9.)  When  the  resulting  radiation  is  into  a  large  solid  angle  the  radiation  resistance, 
or  real  part  of  the  fluid  impedance,  is  low  when  the  length  of  the  radiated  wave  substan- 
tially exceeds  the  diameter  of  the  radiator.  The  diaphragm  serves  to  couple  the  air,  which 
has  low  impedance  per  unit  area,  to  a  motor  having  a  relatively  high  mechanical  impedance, 
•when  connected  to  its  source  of  electrical  energy,  viewed  from  the  diaphragm. 

The  low  radiation  resistance  is  unfortunate  both  because  of  the  problem  of  obtaining 
efficient  energy  transfer  and  because  large  diaphragm  amplitudes  are  required  to  radiate 
appreciable  power  at  low  frequencies.  Figure  1  shows  the  peak  diaphragm  amplitude,  or 
half  the  total  diaphragm  displacement,  required  of  several  piston  sizes  to  radiate  1  watt. 
One-tenth  this  amplitude  is  required  to  radiate  10  milliwatts.  These  curves  cover  the 
frequency  range  in  which  the  radiation  resistance  is  proportional  to  the  square  of  the  fre- 
quency and  to  the  fourth  power  of  the  piston  radius. 


ACOUSTIC  RADIATORS 


13-09 


Most  diaphragms  are  conical  in  shape  and,  if  rigid,  displace  the  same  amount  of  air  for 
the  same  amplitude  as  would  a  fiat  piston  having  a  diameter  equal  to  the  diameter  of  the 
base  of  the  cone.  At  low  frequencies  the  base  diameter  of  the  cone  is  therefore  taken 
as  the  equivalent  piston  diameter. 

The  size  of  the  usual  conical  diaphragm,  frequently  called  a  cone,  is  limited  by  the  need 
for  increasing  its  mass  per  unit  area  as  the  area  is  increased  in  order  to  maintain  adequate 
stiffness.  It  is  customary  to  make  the  diaphragm  thickness  vary  almost  directly  with  the 
diaphragm  diameter.  This  largely  offsets  the  improved  resistance-reactance  ratio  of  the 
fluid  impedance. 

The  conical  diaphragm  may  be  thought  of  as  a  conical  transmission  surface  in  which 
there  are  both  radial  and  circumferential  waves.  At  low  frequencies  only  the  radial  wave 
need  be  considered.  The  flexible 
annular  support  or  "surround"  pro- 
vides a  termination  for  the  base  or 
large  diameter  of  the  conical  surface. 
The  flexural  phase  velocity  of  the 
radial  wave  in  the  diaphragm  and 
the  impedance  of  the  termination 
are  such  that  the  effective  length  is 
one-quarter  wave  at  600  to  1000  cps 
in  large  diaphragms  (16-in.  to  10-in. 
"pistons")  and  at  1000  to  2000  cps 
in  small  diaphragms  (8-in.  to  2-in, 
"pistons").  Below  this  frequency 
all  parts  of  the  cone  move  in  phase 
although  the  amplitude  is  not  sub- 
stantially uniform  except  at  lower 
frequencies  where  the  radial  length 
is  a  small  fraction  of  a  wavelength. 
If  the  annular  support  behaves  as  a 
non-linear  stiffness  over  the  required 
amplitude  range,  as  it  frequently 
does,  the  diaphragm  may  flex  in  a 
complicated  way  even  at  low  fre- 
quencies. 

Wave  transmission  in  the  dia- 
phragm is  desirable  because  it  results 
in  a  more  favorable  high-frequency 
driving  point  impedance  into  which 
the  motor  is  coupled.  It  also  results 


.01 


40 


60  SO  IOQ         20O 
Frequency  in  CPS 


40O    600     10OO 


PIG.  1.  Displacement  of  Piston  from  Equilibrium  Position 
Required  to  Radiate  1  Watt.  The  total  displacement  is 
twice  the  value  given.  (If  both  sides  radiate,  divide  by 


in  a  broader  high-frequency  direc- 
tional pattern  than  would  obtain  if  the  diaphragm  were  a  rigid  piston.  When  maximum 
loudness  is  required,  a  diaphragm  made  of  materials  having  low  internal  dissipation  such 
as  pressed  or  calendered  paper  and  a  low-resistance  flexible  annular  termination  is  used. 
When  a  smoother  response-frequency  curve  is  desired  with  reduced  transient  distortion 
and  the  reduced  loudness  can  be  tolerated,  a  "soft"  more  blotterlike  material  with  higher 
fiexural  resistance  is  employed,  sometimes  with  a  dissipative  termination  of  leather,  felt, 
cloth,  or  a  dissipative  elastomer.  The  modes  of  vibration  of  the  diaphragm  are  also  in- 
fluenced by  cone  angle,  lumped  masses  and  compliances  (usually  annular  beads  or  corruga- 
tions), impregnants,  etc.  Because  of  the  complex  behavior  of  diaphragms  most  design 
work  is  largely  empirical. 

To  reduce  the  mass  per  unit  area  and  yet  obtain  the  benefit  of  a  large  radiating  area 
multiple  diaphragms  are  frequently  used.  If  all  the  diaphragms  vibrate  in  phase  and  with 
the  same  amplitude  the  average  radiation  impedance  seen  by  the  array  will  correspond 
to  that  of  a  single  diaphragm  of  equal  area.  (See  p.  5-66.)  By  properly  orienting  the 
speakers  the  spatial  radiation  pattern,  at  high  frequencies,  may  be  improved. 

In  addition  to  the  common  right  circular  cone  shape  two  others  are  sometimes  used. 
The  conoidal  or  "curvilinear"  is  used  when  an  increase  in  radiation  above  5000  cps  is 
wanted  at  the  expense  of  the  2000-5000  cycle  region.  The  elliptical  shape  is  sometimes 
used  when  space  requirements  limit  one  dimension  of  the  cone.  Unfortunately  the  limited 
dimension  is  usually  the  vertical  one,  resulting  in  the  major  or  long  axis  of  the  ellipse  being 
mounted  horizontally.  Contrary  to  popular  belief  this  leads  to  a  narrow  horizontal  and 
wide  vertical  high-frequency  directional  pattern  as  predicted  from  theoretical  considera- 
tions. In  spite  of  the  appeal  of  its  shape  the  oval  or  elliptical  diaphragm  has  had  limited 
acceptance  because  it  is  more  difficult  to  fabricate,  its  response  is  more  difficult  to  adjust, 


13-10  ELECTROMECHANICAL-ACOUSTIC  DEVICES 

it  has  less  radiation  resistance  in  the  middle-frequency  range  than  a  circular  diaphragm 
of  identical  area,  and  its  asymmetrical  support  leads  to  cone  and  moving-coil  distortion 
which  necessitate  larger  air-gap  clearances  for  a  comparable  safety  factor. 

Over  the  frequency  range  in"  which  the  fluid  mass,  m/,  is  substantially  constant  most 
conical  diaphragms  behave  as  a  rigid  piston  of  mass  mm.  If  the  flexible  centering  members 
supporting  each  end  of  a  conical  diaphragm  are  linear  (displacement  proportional  to  force) , 
their  stiffnesses  may  be  combined  into  an  equivalent  stiffness  sm  or  equivalent  com- 
pliance Cm. 

HORNS.  The  radiation  from  a  horn  may  be  considered  as  originating  at  the  mouth, 
with  the  remainder  of  the  horn  serving  to  "match"  the  mouth  and  throat  terminating 
impedances.  Tor  this  purpose  it  is  important  that  the  walls  be  relatively  non-porous  and 
non-vibratile. 

The  theory  has  been  developed  on  the  basis  of  a  straight  axis  horn  of  circular  crps*- 
section,  with  no  reflections  from  the  mouth.  One  common  departure  from  these  idealized 
conditions  is  in  the  shape  of  the  axis.  It  may  be  bent,  as  in  the  low-frequency  horns  used 
in  theater  systems,  or  it  may  be  folded,  as  in  re-entrant  public-address  horns.  The  contour 
of  the  boundary  at  the  changes  in  the  direction  of  the  axis  must  usually  be  selected  em- 
pirically because  of  the  complex  manner  in  which  the  wave  front  executes  the  change  in 
direction.  The  contour  must  also  be  selected  so  as  to  avoid  exciting  transverse  modes  of 
vibration  of  the  air.  This  requires  smoothly  and  symmetrically  changing  contours,  and 
careful  attention  to  the  symmetry  of  the  driving  diaphragms  with  respect  to  the  throat 
of  the  horn. 

In  practice,  the  cross-sectional  area  may  be  circular,  rectangular,  or  annular,  the  last 
two  corresponding  to  the  theater  and  public-address  horns  mentioned  above.  In  these 
horns  the  lateral  dimensions  are  chosen  to  retain  the  same  area-axial  distance  relation  as 
in  the  circular  (reference)  horn. 

The  effect  of  mouth  reflections  may  be  minimized  by  proper  attention  to  the  horn 
driver  unit  (see  article  3) .  Advantage  may  be  taken  of  reflections  to  load  the  horn  unit 
at  frequencies  near  cutoff  for  which  the  radiation  resistance  is  normally  low. 

5.  EFFICIENCY 

The  total  power  dissipated  by  the  mechanical  system  is 


Pmf  =   S*Tmf  -    &(Tm  +  r,)    -   — f-  (Tm  +  Tf)  (1) 

in  which  rm  and  r/  are  the  mechanical  and  fluid  resistances,  /  is  the  current  in  the  electric 
mesh  and  2*2  is  the  total  self-impedance  of  the  mechanical  system,  and  the  electrical  and 
mechanical  meshes  are  numbered  1  and  2  respectively.  (See  Section  5,  article  33  and 
eqs.  [4a]  to  5J>]  below.)  - 

Energy  Efficiency.  The  ratio  of  acoustic  power  (or  energy)  output  to  electric  power  (or 
energy)  input  is  called  the  energy  or  conversion  efficiency  and  corresponds  to  the  definition 
of  efficiency  commonly  used  for  most  transducers  other  than  electroacoustic  ones.  This 
efficiency  is 


where  rmc  is  the  blocked  resistance  of  the  system. 

System  Efficiency.  Because  both  the  modulus  and  phase  angle  of  the  normal  input 
impedance  of  the  speaker  vary  with  frequency,  the  speaker  in  general  absorbs  less  power 
from  the  source  than  an  ideal  load  or  transducer  would.  Since  this  inability  of  the  trans- 
ducer to  absorb  maximum  power  limits  the  useful  power  output  of  the  source,  it  in  effect 
reduces  the  "efficiency"  of  the  transducer. 

To  take  this  property  into  account,  the  ratio  of  the  acoustic  power  output  to  the  electric 
power  input  which  the  source  would  supply  if  connected  to  an  ideal  transducer  is  defined 
as  the  system  or  absolute  efficiency.  If  the  source  is  a  vacuum  tube  or  generator  whose 
internal  impedance  is  a  "pure  resistance,  Tg,  it  supplies  maximum  power  to  a  resistance  of 
equal  value,  and  the  system  efficiency  is 

•*+*ffiw  (3) 

where  ZM  —  zi2*/z&,  the  other  quantities  are  as  defined  above,  and  the  vertical  lines  indicate 
that  the  absolute  value  is  to  be  taken. 


MOVING-CONDUCTOR  SPEAKERS 


13-11 


6.  MOVING-CONDUCTOR  SPEAKERS 

"A  magnetic  speaker  is  a  loud  speaker  in  which  the  mechanical  forces  result  from  mag- 
netic reactions." 

"A  moving-conductor  speaker  is  a  magnetic  speaker  in  which  the  mechanical  forces  result 
from  magnetic  reactions  between  the  field  of  the  moving  conductor  and  the  steady  applied 
field.  (This  is  sometimes  called  a  dynamic  speaker.)  IT  This  classification  includes  moving- 
coil  or  electrodynamic  (sometimes  called  "dynamic")  and  ribbon  or  "band"  speakers. 
Cross-sections  of  two  typical  mov- 
ing-coil speakers  are  shown  in  Fig.  2.  raffle  Gasket 

The  mechanical  circuit  may  be  L_jAnnuiu«__  jCone  T0ust  Cap 
considered  a  series  circuit  with  the 
elements  m™/,  Cm/,  and  rm/,  which 
include  the  fluid  impedance.  In  gen- 
eral the  circuit  elements  are  func- 
tions of  frequency. 

The  actuating  force  on  the  con- 
ductor is  jSK  dynes,  where  &  is  the 
flux  density  in  gausses,  I  the  con- 
ductor length  in  centimeters,  and  i 
the  instantaneous  current  in  ab- 
amperes  (amperes  X  10).  If  the 
electrical  circuit,  including  the  mov- 
ing coil  and  generator,  has  a  total 
series  inductance  Le,  capacitance  C«, 
and  resistance  re,  and  the  generator 
an  instantaneous  open-circuit  volt- 
age e,  the  instantaneous  "force" 
equations  are 


Cone  Housing 


Field  Coil  Case 
Field  Coll 
•Pole  Piece 


+  req  +  77  4- 

v-<« 


(4a) 


7r--#$  =  0     (46) 


Piece 
(6) 
FIG.  2.    Moving-coil  Speakers,     (a)  Direct  radiator  type; 


(6)  horn  type. 


-Magnet  1— Pole  Piece  i— Wool  Filling 

The  further  assumption  is  made  in 
these  equations  that  (3  is  a  constant, 
independent  of  s,  that  q  does  not 
alter  the  magnetic  field,  and  that 
there  is  negligible  mutual  impedance  between  the  moving  coil  and  the  "field"  coil  which 
provides  the  static  field.  If  j3  is  supplied  by  a  permanent  magnet,  a  constant  fiux  source  is 
approximated  which  is  little  altered  by  q.  The  performance  of  moving-coil  speakers  de- 
pends on  j8  and  not  on  whether  this  is  supplied  by  a  permanent  magnet  or  an  electromagnet. 
If  e  =  E  sin  coi,  the  steady-state  solution  of  eqs.  (4)  is 


E  =  2l2s  -4-  Ziil 


(So) 


0  =  222S  —   Zi$I  (56) 

where  E,  I,  and  s  are  the  rms  steady-state  voltage,  current,  and  velocity,  and  z&  =  fit  is 
the  force  factor.  "The  force  factor  of  an  electroacoustic  transducer  is  a  measure  of  the 
coupling  between  its  electrical  and  mechanical  systems.  It  is  the  ratio  of  the  open-circuit 
force  or  voltage  in  the  secondary  system  to  the  current  or  velocity  in  the  primary  system." 
Equations  (5)  give 


-i  (65) 

f 

DIRECT  RADIATOR  OR  HORNLESS  SPEAKER.  These  are  the  commonly  used 
electrodynamic  speakers  which  are  designed  to  radiate  as  efficiently  as  possible  into  a  solid 
angle  of  the  order  of  27r(' 'semi-infinite  medium").  They  are  used  in  baffles  or  cabinets 
when  response  at  low  frequencies  is  required  but  space  limitations  prevent  the  use  of  a 
horn  or  "directional  bafHe"  to  improve  the  etficiency.  The  radiated  sound  energy  is 
s2?-/,  where  s  is  given  in  eq.  (65)  and  r/  is  SrA,  with  r±  from  Fig.  1,  p.  13-03,  provided  the 


13-12 


ELECTROMECHANICAL-ACOUSTIC  DEVICES 


mounting  approximates  an  infinite  baffle  at  the  frequency  considered.  In  the  low  and 
low  middle  frequencies,  the  fluid  resistance  is  proportional  to  the  square  of  the  frequency. 
Therefore,  if  SV/  is  to  be  approximately  constant,  the  velocity  s  must  vary  inversely  with 
frequency.  From  eq.  (66)  it  may  be  seen  that,  when  the  applied  frequency  is  appreciably 
higher  than  the  resonant  frequency  of  the  mechanical  system,  s  is  largely  a  function  of  «12 
and  222-  Since  212  is  approximately  a  constant,  and  222  ==  &#»»»/,  *  varies  inversely  with  w 
particularly  if  212  is  small.  The  radiated  acoustic  power  is  therefore  approximately  in- 
dependent of  frequency.  In  this  frequency  range  the  system  efficiency,  r)8,  is  approximately 


2  X 


(7) 


where  a.  is  the  ratio  of  actual  conductor  volume  to  air-gap  volume,  V  (product  of  gap 
length,  mean  perimeter,  and  coil  winding  length)  ; 


E  =  —  —  ==  gap  energy  (ergs) 

07T 

B  is  the  average  flux  density  in  volume  V  (gauss)  ;  C  is  the  conductivity  of  the  conductor 
with  respect  to  copper;  m  is  the  effective  motor  mass  (voice  coil,  cone,  fluid  mass)  (gram)  ; 
and  d  is  the  effective  piston  diameter  (inches)  . 

When  the  velocity  of  the  diaphragm  is  largely  limited  by  the  mass  reactance  of  the 
mechanical  system  and  varies  (roughly)  inversely  with  frequency,  the  device  is  said  to 
have  ''inertia  control."  To  obtain  uniform  response,  the  natural  frequency  is  placed  near 
the  lowest  frequency  to  be  transmitted.  In  12-in.  speakers  this  resonant  frequency  is  of 
the  order  of  75  cps.  At  very  low  frequencies,  either  the  stiffness  reactance  or  the  mechan- 
ical resistance  limits  the  velocity,  and  the  response  decreases  even  if  the  speaker  is  operated 
in  an  infinite  bafiie.  In  the  middle  frequency  range,  the  normal  impedance  and,  therefore, 
the  force  on  the  mechanical  circuit  are  fairly  constant.  In  general  the  damping  is  less 
than  "critical"  or  the  value  required  to  give  the  shortest  transient  response.  See  "Tran- 
sient Response,"  Section  5,  articles  13  and  15. 

Figure  3  indicates  the  result  of  varying  /3Z  (or  212)  on  the  pressure  measured  in  the  sound 
field.  Since  the  speaker  directivity  is  almost  independent  of  frequency  in  this  range,  the 
ordinates  are  proportional  to  the  total  radiated  acoustic  power.  These  curves  show  that 


~s 

JO. 

:^r 

"^ 

£ 

Q      10 

v" 

"N.^ 

v/ 

^ 

V 

-»^. 

12000 

S 

/. 

// 

-•x 

X**r 

v 

•. 

u       8500 

0. 

$ 

^ 

^ 

\ 

\ 

•'-. 

... 

6000 

5 

--?<] 

*  / 

>- 

->-, 

"•^. 

4250 

"« 

"•*".•** 

/ 

C£ 

Of) 

/ 

/ 

40 


50 


60  70         80        90     100 

Frequency  in  Cycles  per  Second 


150 


FIG.  3.     Variation  in  Acoustic  Pressure  with  Flux  Density,  or  Force  Factor.     (Measurements  were 
made  on  the  axis,  10  feet  from  a  direct  radiator  moving  conductor  speaker.) 

speakers  having  large  force  factors  or  values  of  212  and  high  efficiency  in  the  middle  fre- 
quency range  have  relatively  less  response  near  the  resonant  frequency  of  the  mechanical 
circuit.  Increasing  the  flux  density  and  force  factor  increases  the  energy  efficiency  but 
decreases  the  system  efficiency  near  resonance.  Because  the  Tna.yTrn.riTn  peak  powers  in 
speech  and  music  (see  Section  12,  articles  10  and  11)  occur  in  the  middle  frequency  range, 
it  is  important  that  the  speaker  have  high  efficiency  in  this  range,  and  the  better  speakers 
have  high  force  factors,  which  also  reduce  the  transient  distortion  of  the  speaker. 


MOVING-CONDUCTOR  SPEAKERS 


13-13 


Direct  radiator  speakers  normally  reach  their  maximum  efficiency  in  the  low  middle 
frequency  range,  except  when  the  effective  baffle  size  is  large  so  that  the  resonant  fre- 
quency is  radiated  effectively.  In  that  case,  if  z&  is  small  the  maximum  system  efficiency 
may  occur  at  the  frequency  of  mechanical  resonance.  The  normal  impedance  is  resistive, 
or  largely  so,  in  either  case.  (See  Fig.  4.) 


+  80 


40      60  80  100        2OO          400    600     lOOO 
Frequency  Jn  Cycles  per  Second 


2000        4000 


10,000 


FIG.  4.     Blocked  and  Normal  (Primed)  Resistance  and  Reactance  of  Moving-coil  Direct^radiator 

Speaker  in  Ohms 

The  system  efficiencies  hi  the  middle  frequency  range  vary  from  roughly  1  to  20  per 
cent  in  commercial  direct  radiator  speakers.  "The  energy  efficiency  values  are  slightly 
higher  in  the  middle  frequency  range  and  exceed  90  per  cent  at  the  mechanical  resonant 
frequency  in  some  designs. 

ENCLOSURES.  Direct  radiator  speakers  are  often  placed  in  enclosures  to  contain 
and  control  the  radiation  from  the  rear  surface  of  the  diaphragm.  If  the  enclosure  is 
complete,  that  is,  has  no  vents,  it  is  sometimes  called  a  "total  enclosure"  (see  Enclosed 
Back  Piston,  article  1).  The  rear  radiation  may  be  put  to  use  by  permitting  it  to  escape 
from  the  enclosure  after  first  modifying  it  in  phase  and  magnitude  by  a  suitable  acoustic 
network.  When  the  vents,  or  ports,  which  radiate  the  energy  from  the  rear  of  the_  di- 
aphragm are  close  to  the  front  surface  of  the  diaphragm,  the  mutual  impedance  isjhigh. 
In  this  case  the  phase  of  the  rear  radiation  is  critical  but  gives  rise  to  maximum  radiation 
when  it  is  properly  controlled  (see  article  2) . 

When  the  length  of  the  radiated  wave  is  large  compared  with  the  enclosure  dimensions 
the  acoustic  parameters  may  be  considered  "lumped."  A  single  cavity  enclosure  with  a 
port  may  then  be  considered  to  add  a  compliance  corresponding  to  that  seen  by  the 
diaphragm  with  the  port  blocked  or  covered,  which  is  in  series  with  the  speaker  compliance. 
The  enclosure  compliance  is  shunted  by  the  effective  mass  and  radiation  resistance  of  the 
port.  The  latter  values  are  those  referred  to  the  speaker  diaphragm.  The  frequency  of 
the  two  resulting  low-frequency  modes  may  be  computed  very  approximately  by  neglect- 
ing the  mutual  reactance  term  arising  from  the  coupling  of  the  external  radiating  surfaces 
of  the  diaphragm  and  port  through  the  mutual  fluid  impedance.  The  real  part  of  this 
mutual  impedance  must,  however,  be  included  in  any  accurate  calculation  of  the  total 
radiated  power.  .  .  .^ 

The  vented  enclosure  may  be  used  to  maintain  uniform  radiation  down  to  lower  fre- 
quencies than  those  radiated  effectively  by  a  total  enclosure  of  identical  volume,  or  it 
may  be  used  to  provide  a  rise  in  the  radiation  near  the  cutoff  frequency  of  the  mvented 
enclosure.  In  practice  it  is  common  to  provide  a  compromise  between  these  two  by  so 
choosing  the  speaker  and  enclosure  parameters  that  the  resulting  two  modes  of  the  simple 
structure  described  above  occur  approximately  one-half  octave  below  and  above  the  mode 
of  the  speaker  and  enclosure  with  blocked  port.  The  augmented  low-frequency  radmtiom 
is  obtained  with  substantially  reduced  non-linear  distortion  since  most  of  the  radiation  is 
from  the  port,  which  need  have  no  variable  or  non-linear  parameters,  and  the  diaphragm 
displacement  is  appreciably  reduced.  » ,  ,  ,  ,  -,.  *  i 

The  rear  of  the  diaphragm  is  sometimes  coupled  to  a  highly  absorbent  kne  approximately 
one-quarter  wave  long  near  the  cutoff  frequency  of  the  acoustical  system.  The  hne  ab- 
sorption should  be  low  in  the  frequency  interval  in  which  the  line  is  between  three-quarters 
and  one  wave  long  and  increase  rapidly  above  the  upper  frequency  of  this  interval  to 
suppress  radiation  which  would  otherwise  be  out  of  phase  with  the  front  radiation.  When 


13-14 


ELECTROMECHANICAL-ACOUSTIC  DEVICES 


the  line  is  one-quarter  and  three-quarter  wave  long  it  serves  as  an  impedance  inverter, 
thereby  raising  the  impedance  seen  by  the  diaphragm  and  reducing  its  amplitude  for  a 
given  total  radiation. 

HORN-TYPE  MOVING-COIL  (OR  CONDUCTOR)  SPEAKERS.  To  increase  the 
fluid  resistance,  r/,  a  horn  may  be  used  to  couple  the  diaphragm  to  the  acoustic  medium. 
Because  of  their  greater  throat  resistance  at  low  frequencies  for  a  given  size,  hyperbolic- 
exponential  horns  with  T  <  1  (see  article  3)  are  normally  used. 

In  the  case  of  horn  units  the  normal  (electrical)  impedance  can  be  made  quite  uniform 
(see  Fig.  5)  so  that  the  force  on  the  diaphragm  is  approximately  constant  if  the  source 
voltage  and  impedance  are  constant. 


1000 
800 
600 

400 


200 


100 
80 


40 


W  20 


10 

8 

6 


\ 


\ 


r 


&/ 


30  40     60  80100 


~&y» 


200          400    600     1000        2000 
Frequency  in  Cycles  per  Second 


40006000    10000 


FIG.  5.    Relative  Scalar  Normal  Impedances  of  Typical  Loud  Speakers.     (All  arbitrarily  adjusted  to 
have  the  same  impedance  at  400  cps.    Reactance  of  an  inductance  and  capacitance  plotted  for  com- 
parison.) 

The  diaphragm  is  made  to  resonate  an  octave  or  two  above  the  horn  cutoff  frequency. 
The  stiffness  reactance  of  the  diaphragm  assembly  then  reduces  the  effect  of  the  mass 
reactance  of  the  horn  near  its  cutoff  frequency.  At  high  frequencies,  the  stiffness  of  the 
sound  chamber  may  be  made  to  reduce  the  effect  of  the  mass  reactance  of  the  diaphragm. 
The  fluid  resistance  is  adjusted  to  the  desired  value  by  proper  choice  of  the  ratio  (£<;/£*). 
These  factors  make  possible  the  design  of  a  horn  having  uniform  response  over  an  extended 
frequency  range. 

At  low  frequencies,  irregularities  in  response  result  from  the  variation  in  throat  imped- 
ance of  a  horn  of  finite  length.  The  resonant  frequencies  of  the  horn  near  its  cutoff  fre- 
quency also  give  rise  to  transient  distortion.  By  using  a  horn  unit  having  a  large  force 


MAGNETIC-AEMATUEE  SPEAKERS 


13-15 


factor  and  by  proper  choice  of  the  other  elements,  the  steady-state  variation  in  response 
and  the  transient  distortion  may  be  made  negligible. 

At  high  frequencies  destructive  interference  occurs  in  the  sound  chamber  since  its  dimen- 
sions are  of  the  order  of  magnitude  of  the  length  of  the  sound  wave.  The  horn  throat  is 
therefore  sometimes  so  constructed  that  the  difference  in  path  length  from  different  parts 
of  the  diaphragm  to  the  throat  is  a  minimum.  The  use  of  a  phase  equalizing  plug  for  this 
purpose  is  shown  in  the  horn  unit  of  Fig.  2.  At  very  high  frequencies  the  diaphragm  no 
longer  behaves  as  a  rigid  piston,  and  this  alters  the  response  of  the  unit. 

The  minimum  compliance  of  the  sound  chamber  is  limited  by  its  minimum  volume,  TV 
This  volume  must  be  adequate  to  provide  ample  mechanical  clearance  at  low  frequencies, 
where  the  diaphragm  amplitude  is  a  maximum,  and  to  limit  the  distortion  which  results 
from  "finite"  sound  pressures. 


Armature 
Extension' 


7.  MAG1TOTIC-ARMATDEE  SPEAKERS 

"A  magnetic-armature  speaker  is  a  magnetic  speaker  whose  operation  involves  the 
vibration  of  the  ferromagnetic  circuit.  (This  is  sometimes  called  an  electromagnetic 
speaker.)"  Numerous  magnetic-armature  designs 
have  been  proposed.  The  principle  of  operation  of 
this  type  of  speaker  is  analogous  to  that  of  moving- 
conductor  speakers,  and  equations  for  the  latter 
apply  to  this  type  if  the  appropriate  value  of  zu  is 
used.  They  differ  principally  in  the  fact  that  the 
conductor  does  not  move,  which  permits  the  use  of  a 
large  conductor  volume.  In  practice,  high  loudness 
efficiencies  can  be  obtained  in  a  limited  frequency 
range  with  a  magnet  having  moderate  magnetomo- 
tive force.  The  efficiency  at  very  high  frequencies 
is  normally  poor  because  of  losses  in  the  armature. 
The  very  low-frequency  response  is  limited  by  the 
stiffness  required  to  give  adequate  armature  sta- 
bility. The  current  displacement  curve  is  linear 
over  a  very  limited  range  and  gives  rise  to  appreci- 
able amplitude  distortion. 

BALANCED-ARMATURE  MAGNETIC 
SPEAKER.  A  cross-sectional  view  of  a  unit  of  this 
type  is  shown  in  Fig.  6.  The  speech  current  flows 
through  the  stationary  coil  which  surrounds  the 
armature.  It  increases  the  effective  flux  through  two 
(diagonally  located)  gaps  and  decreases  it  in  the 
other  two  gaps.  The  steady  flux,  <£>o,  is  increased 
and  decreased  by  an  amount  <£  =  4acNi/Gt  gausses, 


;  Laminated 
PoSe  Piece 


Fia.  6.    Balanced  Armature  Speaker 


where  N  is  the  number  of  turns  on  the  coil,  i  is  the  current  through  the  coil  in  abamperes, 
and  (Ft  is  the  effective  reluctance  of  the  alternating  flux  path. 

The  total  effective  force  on  the  armature,  acting  at  the  magnetic  force  center,  is 


4-  < 


87T.4 


ffi 


(8) 


where  F  is  the  rms  force  in  dynes  and  A  is  the  effective  area  in  square  centimeters  at  each 
gap  and  jSo  is  the  steady  or  undisturbed  flux  density.  For  small  amplitudes  the  reluctance 
of  the  alternating  flux  path  is  approximately  constant,  since  most  of  the  reluctance  is  in 
the  gap,  and  the  reluctance  of  the  permanent  magnet  path  is  large. 

The  assumption  is  usually  made  that  all  the  elements  of  the  circuits  and  the  force  factor 
are  constant.  This  is  approximated  only  when  the  amplitude  is  small.  When  the  arma- 
ture is  in  the  undisturbed  center  position,  it  is  in  a  state  of  unstable  equilibrium.  The 
application  of  a  force  results  in  a  force  tending  to  increase  the  displacement.  This  property 
is  called  negative  stiffness  or  compliance.  (See  Section  5,  article  30.)  It  here  results  from 
the  fact  that  the  torque  on  the  armature  is  proportional  to  the  square  of  the  flux  density. 
Therefore,  it  increases  more  rapidly  at  the  tip  that  is  approaching  one  magnet  tip  than 
it  decreases  at  the  tip  that  is  receding  from  the  other  magnet  tip.  Since  the  force  varies 
with  the  square  of  the  flux  density,  and  the  density  varies  almost  inversely  with  distance 
between  the  armature  and  magnet  tip,  the  stiffness  can  be  considered  a  constant  only  for 
small  displacements.  The  positive  value  of  Cm  is,  therefore,  the  difference  between  the 
positive  diaphragm  (and  fluid,  if  any)  compliance  and  the  negative  compliance.  Stability 


13-16  ELECTBOMECHANICAL-ACOUSTIC  DEVICES 


of  the  system  requires  that  the  rate  of  change  of  force  with  displacement  be  less  than  the 
coefficient  of  stiffness.  The  further  assumption  is  made  that  the  lever  arm  has  infinite 
stiffness.  In  practice,  resonances  of  this  arm  play  a  large  part  in  modifying  the  mechanical 
impedance  at  high  frequencies. 

The  steady-state  solution  for  this  speaker  is  the  same  as  that  of  eqs.  (5)  and  (b;.  In 
this  case  zn  =  4faN/(R;  for  the  more  exact  theory  012  is  considered  complex. 

The  principle  of  operation  is  analogous  to  that  of  the  moving-conductor  speaker.  Equa- 
tions for  the  latter  apply  if  the  above  value  of  z\i  is  substituted.  A  large  conductor  volume 
is  normally  used  to  increase  the  force  factor.  This  makes  the  inductance  of  the  electrical 
circuit  much  higher  than  in  moving-con  speakers.  For  this  reason,  the  normal  impedance 
varies  over  wide  limits.  The  normal  impedance  is  proportional  to  frequency  over  much 
of  the  frequency  range,  as  shown  in  Fig.  5.  The  rise  in  impedance  above  the  value  pre- 
dicted on  this  basis  at  the  high-frequency  end,  in  this  particular  speaker,  was  due  to 
electrical  resonance.  The  speaker  was  of  the  high-impedance  type,  and  the  inductance 
and  distributed  capacitance  resonated  at  the  high-frequency  end.  This  is  normally  the 
case  in  high-impedance  speakers  of  this  type,  and  its  effect  must  be  included  in  any  com- 
plete performance  analysis. 

BIPOLAR  MAGNETIC-ARMATURE  SPEAKER.  This  type  of  speaker  consists  of 
a  steel  diaphragm  mounted  near  the  two  ends  of  a  U  magnet.  The  speech  current  flows 
through  the  stationary  coils  which  surround  the  two  pole  pieces.  If  the  current  through 
the  two  coils,  connected  in  series,  is  /  sin  cot,  and  the  coils  have  N  turns,  the  alternating 

flux  is  <£  =  4         sm  ^  ,  where  I  is  the  maximum  value  of  the  current  in  abamperes,  and 

(R 
(R  is  the  effective  reluctance  of  the  alternating  flux  path. 

The  total  effective  force  F  acting  at  the  magnetic  force  center  is 


~ 


(HA 


(R2A 


(R2A 


^  } 


The  first  term  on  the  right  is  a  constant  force;  the  second  is  the  useful  component  which  is 
proportional  to  <fr>  and  to  the  signal  current.  The  last  two  terms  are  distortion  terms  and 
are  minimized  by  making  <£o  »  <£. 

The  force  factor  212  =  2faN/(R.  The  more  exact  theory  includes  the  case  in  which  212 
is  complex. 

The  principle  of  operation  is  analogous  to  that  of  the  balanced-armature  magnetic  and 
moving-conductor  speakers.  The  equations  for  these  speakers  apply  to  the  bipolar 
magnetic  armature  if  the  value  of  212  given  above  is  used. 


8.  CONDENSER  SPEAKERS 


"A  condenser  speaker  is  a  loud  speaker  in  which  the  mechanical  forces  result  from  electro- 
static reactions."  Speakers  of  this  type  have  a  movable  conducting  electrode  which  serves 
as  the  diaphragm  and  is  mounted  close  to  a  perforated  fixed  electrode  or  between  two 
perforated  fixed  electrodes. 

TWO-ELECTRODE  CONDENSER  LOUDSPEAKER  WITH  MECHANICAL  CIR- 
CUIT HAVING  ONE  DEGREE  OF  FREEDOM.  One  speaker  of  this  type  has  a  movable 
and  a  fixed  electrode  separated  by  a  thin  dielectric  as  shown  in  Fig.  7.  The  force  of  attrac- 

tion between  the  electrodes  as  the  charge 
varies  provides  the  actuating  force  for  the 
movable  electrode  which  is  used  as  the 
diaphragm. 

The  equations  for  a  system  of  this  type 
are  developed  in  Section  5,  article  32.  The 
steady-state  equations  for  a  single  sine-wave 


"  Metal  Leaf  Diaphragm 


Rough  Punched  Holes 
(Not  Burred) 

FIG.  7.     Condenser  Speaker 


applied  voltage  are: 


0   = 


+ 

+ 


(100) 

(10b) 


where  212  =  —  ^rr  =  —7  •    The  force  factor  therefore  varies  directly  with  the  polarizing 
CtfCo»o        fcjcfo 

voltage,  EQ,  and  inversely  with  the  frequency  and  no-signal  separation  of  the  electrodes,  do- 
The  minimum  value  of  the  separation  is  determined  by  the  maximum  low-frequency 


TELEPHONE  RECEIVERS   (EARPHONES)  13-17 

amplitude,  and  it  must  be  large  in  comparison  with  the  amplitude  if  non-linear  distortion 
is  to  be  avoided. 


The  radiated  acoustic  power  is  &r/  X  10~7  watt,  where  *  is  given  by  eq.  (116)  and  rf  by 
article  2.  The  diaphragm  does  not  move  as  a  piston.  Therefore,  a  generalized  velocity 
based  on  the  type  of  deformation  here  obtained  must  be  used. 

Because  the  blocked  impedance  of  the  speaker  is  that  of  a  capacitance,  G,  the  normal 
impedance  varies  inversely  with  frequency  over  much  of  the  frequency  range.  Therefore, 
uniform  response  and  high  efficiency  are  difficult  to  obtain.  The  impedance  of  the  elec- 
trical circuit  is  sometimes  altered  to  improve  the  response. 

9.  PNEUMATIC  SPEAKERS 

"A  pneumatic  speaker  is  a  loudspeaker  in  which  the  acoustic  output  results  from  varia- 
tions of  an  air  stream."  It  is  of  the  irreversible  or  relay  type.  The  definition  of  efficiency 
for  reversible  speakers  is  not  used  in  this  case,  since  the  efficiency,  as  defined  for  them,  can 
exceed  100  per  cent  because  no  account  is  taken  of  the  power  used  to  compress  the  air. 
Any  type  of  loudspeaker  motor  may  be  used  to  drive  a  very  light  balanced  valve  which 
modulates  an  air  stream. 

These  units  permit  the  generation  of  large  acoustic  powers.  The  valve  and  valve  parts 
give  rise  to  spurious  noises  that  are  difficult  to  eliminate.  When  large  acoustic  outputs 
are  generated,  there  is  appreciable  "finite"  amplitude  distortion  in  the  types  that  have 
been  made. 

10.  TELEPHONE  RECEIVERS  (EARPHONES) 

Telephone  receivers  are  so  constructed  that  they  operate  directly  into  the  ear  cavity. 
The  fluid  impedance  of  the  ear  cavity  varies  appreciably  from  ear  to  ear,  particularly  at 
high  frequencies.  If  there  is  no  fluid  leak  between  the  ear  "cap"  of  the  receiver,  which 
held  in  contact  with  the  ear,  and  the  ear,  the  fluid  impedance  is  approximately  that  of  a 
6-cms  cavity  at  frequencies  in  the  middle  and  lower  frequency  ranges.  At  high  frequencies, 
the  ear  cavity  resonates  and  the  fluid  resistance  of  the  cavity  increases.  If  there  is  a  fluid 
leak  between  the  ear  cap  and  ear,  the  fluid  resistance  of  this  leak  must  be  considered  at 
low  frequencies  where  it  tends  to  reduce  the  pressure  in  the  cavity. 

It  has  been  found  that  a  good  compromise  design  for  most  ears  is  one  in  which  the  dia- 
phragm displacement  is  independent  of  frequency.  At  low  frequencies  this  gives  constant 
pressure  in  the  ear  cavity  if  -there  is  no  fluid  leak.  Since  the  magnitude  of  the  reactance 
of  the  ear  impedance  substantially  exceeds  the  resistance  below  1000  cps,  the  power  sup- 
plied to  the  ear  and  the  efficiency,  as  defined  for  other  transducers,  are  not  satisfactory 
performance  criteria.  The  pressure  squared,  produced  at  the  end  of  a  non-dissipative 
cavity  or  "coupler/'  per  unit  power  available  from  an  ideally  terminated  source,  is  a  com- 
mon performance  criterion.  A  &-cc  coupler  is  used  for  ear  cap  or  external  earphones  and  a 
2-cc  coupler  for  insert  receivers  which  are  placed  in  the  outer  ear  canal  and  thereby  reduce 
the  volume  of  the  cavity  between  the  receiver  and  the  eardrum. 

MOVING-CONDUCTOR  TELEPHONE  RECEIVERS.  Both  moving-coil  and  ribbon 
telephone  receivers  are  in  use.  The  construction  of  one  type  of  moving-coil  receiver  is 
similar  to  that  of  the  microphone  shown  in  Fig.  3,  p.  13-24.  The  values  of  the  circuit 
elements  differ  from  those  used  in  the  microphone,  in  order  to  give  as  nearly  as  possible 
constant  diaphragm  amplitude  at  all  frequencies  with  a  constant-voltage  source. 
t  If  the  mechanical  circuit  consists  only  of  the  stiffness  and  resistance  of  the  diaphragm, 
and  the  combined  mass  of  the  diaphragm  and  moving  coil,  its  "force"  equations  are  given 
by  eqs.  (6).  The  displacement  s  =  *//«,  where  a  is  given  by  eqs.  (6),  Since  z&  is  a  con- 
stant, if  zn  is  made  approximately  constant,  2*2  must  vary  inversely  with  frequency  for 
the  diaphragm  displacement  to  be  independent  of  frequency.  This  requires  that  the 
diaphragm  have  stiffness  reactance  over  the  range  of  uniform  response.  This  is  obtained 
by  placing  the  resonant  frequency  of  the  diaphragm  near  the  maximum  frequency  to  be 
transmitted.  Under  these  conditions  the  normal  impedance  of  the  receiver  is  fairly  uni- 
form below  the  frequencies  near  the  resonant  frequency  of  the  diaphragm.  The  force 
is  therefore  nearly  constant.  The  displacement  of  the  diaphragm  is  small,  and  hence  the 


13-18  ELECTROMECHANICAL-ACOUSTIC  DEVICES 

efficiency  of  the  receiver  is  poor  when  the  resonant  frequency  and  stiffness  are  high.  It 
is  also  difficult  in  practice  to  obtain  very  high  resonant  frequencies. 

To  improve  the  efficiency  and  yet  maintain  uniform  response,  additional  mechanical 
circuits  are  sometimes  coupled  to  the  diaphragm.  The  resonant  frequency  of  the  dia- 
phragm itself  is  lowered,  and  the  value  of  the  mechanical  elements  is  chosen  to  give  uni- 
form diaphragm  displacement. 

If  the  mechanical  impedance  is  large  in  comparison  with  the  ear  cavity  impedance,  the 
latter  may  be  neglected.  In  general,  it  may  be  neglected  only  for  approximate  calcula- 
tions. The  impedance  of  the  fluid  between  the  diaphragm  and  the  ear  cap  must  also  he 
considered  in  any  complete  analysis. 

MAGNETIC-ARMATURE  TELEPHONE  RECEIVER.  The  theory  of  operation 
corresponds  to  that  of  magnetic-armature  loudspeakers  given  above.  The  first  center 
moving  mode  of  vibration  of  the  magnetic  diaphragm  in  the  early  type  of  telephone 
receiver  occurs  near  1000  cps.  Near  this  frequency,  the  amplitude  is  large  and  the  pressure 
in  the  ear  cavity  is  large.  Below  this  resonance  region,  the  diaphragm  has  approximately 
constant  stiffness  or  compliance,  and  the  amplitude  is  fairly  uniform.  At  frequencies 
above  the  resonant  region,  the  displacement  decreases  and  goes  through  a  series  of 
decreasing  maxima  at  higher  modes  of  vibration.  In  more  recent  types  there  is  a  thin 
film  of  air  at  the  back  of  the  diaphragm  which  is  coupled  to  an  auxiliary  acoustical 
network.  The  parameters  are  chosen  to  give  uniform  pressure  at  the  bottom  of  the  test 
cavity  or  coupler  up  to  3000  cps  in  telephone  hand  sets  and  up  to  4000  cps  in  some 
military  types. 

PIEZOELECTRIC  TELEPHONE  RECEIVERS.  The  high  frequency  at  which  the 
fundamental  mechanical  resonant  frequency  of  a  small  Rochelle  salt  crystal  occurs  permits 
its  use  as  a  motor  or  motor  and  diaphragm  in  a  telephone  receiver.  The  large  variation 
in  normal  impedance  with  frequency  makes  the  response  more  dependent  on  the  source 
impedance  than  it  is  when  the  normal  impedance  is  constant. 

11.  PERFORMANCE  AND  TESTS 

One  criterion  of  the  excellence  of  a  complete  sound-transmission  system  is  its  ability 
to  produce  the  same  space-pressure-time  pattern  at  the  ears  of  the  listener  that  would  be 
experienced  if  he  were  immersed  in  the  sound  field  in  the  region  of  the  original  sound. 
This  can  be  achieved  with  substantially  complete  realism  in  a  binaural  system  in  which 
two  transmission  links  couple  two  properly  mounted  pressure  microphones  to  two  broad- 
band telephone  receivers  placed  on  the  ears  of  the  listener. 

In  practice  there  is  a  growing  tendency  to  apply  a  second  criterion,  namely  the  ability 
of  a  system  to  provide  reproduction  which  is  judged  to  be  pleasant  by  a  jury  selected  on 
the  basis  of  scientific  sampling  of  the  public  or  the  ultimate  listening  group.  This  second 
criterion  has  grown  in  importance  for  a  number  of  reasons.  One  is  that  most  listeners 
prefer  loudspeakers  to  headphones,  thus  introducing  complex  effects  due  to  coupling  the 
loudspeaker  to  the  ears  of  the  listener  via  the  listening  room.  Probably  the  most  important 
reason  for  choosing  pleasantness  as  the  criterion  of  excellence  is  that  radio  and  phonograph 
systems  have  in  large  measure  attained  the  stature  of  an  art  medium  in  their  own  right. 
This  has  led  to  extensive  work  on  the  design  of  new  types  of  studios  and  the  development 
of  new  pickup  techniques,  all  intended  to  please  the  listener  in  his  home. 

Though  the  complete  test  of  a  speaker  is  a  complicated  process,  all  the  objective 
measurements  have  as  their  goal  a  satisfactory  correlation  between  their  results  and  those 
of  properly  conducted  listening  tests.  Thus  the  ear  becomes  the  final  arbiter  of  the  excel- 
lence of  a  sound  system. 

For  discussing  the  aural  performance  it  is  desirable  to  use  terms  descriptive  of  the 
frequency  ranges  and  of  certain  characteristics  of  the  reproduced  sound.  As  a  group  these 
terms  constitute  the  imagery  by  which  sound  systems  are  described.  They  relate  princi- 
pally to  two  important  frequency  regions:  400  to  800  cps,  which  contains  most  of  the  energy 
of  speech  and  music;  and  2000  to  3000  cps,  where  the  energy  components  contribute  most 
strongly  to  the  loudness.  The  first  region  is  located  in  what  most  listeners  classify  as  the 
low  middles.  A  speaker  with  normal  low  middles  is  said  to  have  "body,"  while  an  excess 
is  "muddy,"  and  a  deficiency  sounds  "thin."  When  the  second  range,  in  the  lower  highs, 
is  normal,  reproduced  sound  is  "crisp";  an  excess  introduces  "bite"  and  is  "brilliant";  a 
deficiency  reduces  articulation  and  loudness  and,  if  no  extreme  highs  are  present  to  provide 
"spit,"  may  yield  a  "mellow"  speaker. 

The  whole  range  is  conveniently  divided  into  the  very  low  frequencies,  below  100  cps, 
lows  from  100  to  250  cps,  lower  middle  from  250  to  800  cps,  upper  middles  from  800  to 
2000  cps,  lower  highs  from  2000  to  4400  cps,  middle  highs  from  4400  to  8000  cps,  and 


PERFORMANCE  AND  TESTS 


13-19 


extreme  highs  above  8000  cps.  The  boundaries  are  quite  arbitrary  and  win  depend  on 
the  auditor,  listening  environment,  and  program  material. 

>  OBJECTIVE  PERFORMANCE  CRITERIA.  Loudspeaker  performance  is  determined 
to  a  large  extent  by  the  response,  efficiency,  directivity,  distortion,  power  capacity,  and 
impedance.  The  relative  importance  of  these  and  other  less  important  performance 
criteria  depends  on  the  application  involved. 

The  response  is  that  characteristic  of  the  acoustic  output,  expressed  as  a  function,  of 
frequency,  to  which  a  particular  aural  effect  is  closely  proportional.  The  type  of  response 
will  depend  on  the  use  to  which  the  speaker  is  put.  For  speakers  used  outdoors  or  for  very 
directional  speakers  the  free  space  axial  sound  pressure  level,  as  a  function  of  frequency, 
is  used  and  is  called  the  sound  pressure  response. 

For  use  indoors  the  sound  pressure  level  is  determined  by  the  total  acoustic  power 
radiated  and  by  the  shape,  size,  and  acoustical  impedance  of  the  room  and  its  contents. 
Thus  the  effect  of  the  speaker  alone  is  approximated  by  the  total  radiation,  usually  ascer- 
tained by  integration  of  the  sound  pressure  measured  in  free  space  at  fixed  radius  vector 
and  variable  angle.  The  total  radiation  response  is  modified  by  the  "room  response"  to 
produce  the  desired  effect  on  the  ear,  and  it  is  often  helpful  to  measure  the  rms  average 
sound  pressure  in  a  room  which  is  representative  for  the  intended  applications  of  the 
speaker.  This  mean  pressure  (corresponding  to  the  mean  energy  density)  response  has 
good  correlation  with  the  aural  effect  when  the  measuring  room  is  also  that  used  for 
listening. 

In  Fig.  8  are  shown  the  sound  pressure,  total  radiation,  and  mean  energy  density  re- 
sponse curves  for  a  direct  radiator  speaker  in  a  finite  baffle.  The  excess  of  high  frequencies 


+10 


+ 

01 


»   —5 

8 

u 

c 

5-15 
I 
|-20 

1 
2~~ 

-30 
-35 


-40 


Axial  Sound  Pressure,  Free  Space 

_ Total  Radiation,  Dead  Room 

- — - — -  Mean  Energy  Density,  Average  Room 


40       60   80  100 


200 


40OO  60OO   10000        20000 


400     600       10OO          2OOO 
Frequency  in  Cycles  per  Second 
FIG,  8.    Three  Types  of  Response-frequency  Curves  for  a  Direct  Radiator  Speaker  in  a  Finite  Baffle 

in  the  pressure  response  is  due  to  directionality,  and  the  difference  between  the  second 
and  third  is  due  to  the  "response"  of  the  room. 

The  system  efficiency  of  speakers  has  been  discussed  in  article  5.  The  loudness  effi- 
ciency of  two  speakers  is  compared  using  a  generator  of  constant  emf  and  impedance, 
attenuation  being  inserted  in  the  amplifier  for  the  louder  speaker.  This  judgment  is 
usually  the  first  of  any  listening  test.  It  is  becoming  standard  practice  to  test  speakers 
with  a  source  whose  internal  impedance  is  equal  to  its  rated  load  impedance.  The  basis 
of  stating  the  input  to  the  speaker  is  conveniently  the  maximum  power  that  may  be  taken 
from  such  a  source,  and  it  is  called  the  power  available  to  the  speaker. 

The  series  of  sound  pressure-frequency  response  curves  run  at  various  azimuths  depict 
the  directional  characteristics  of  the  speaker.  Another  clue  to  the  directivity  is  obtained 
by  comparing  the  sound  pressure  and  total  radiation  response  curves  (see  Fig.  8).  When 
the  wavelength  is  long  compared  to  the  size  of  the  radiator,  the  radiation  is  uniform,  and 
th§  curves  should  coincide.  The  differences  at  higher  frequencies  then  indicate  the  magni- 
tude of  the  directional  effects. 

If  the  listener's  ear  is  relied  on  completely  to  select  speakers  having  high  loudness  and 
(apparent)  bass  and  treble  response,  there  usually  result  speakers  having  large  amounts  of 


13-20  ELECTROMECHANICAL-ACOUSTIC  DEVICES 


800 


600 


400 


300 


non-linear,  transient,  and  frequency  response  distortion.    The  need  for  quantitative  tests 

is  indicated  by  Fig.  9,  which  shows 
the  non-linear  distortion  in  the  sound 
pressure  of  a  typical  12-in.  speaker 
mounted  in  an  8  by  8  ft  baffle.  The 
speaker  has  an  equivalent  piston 
diameter  of  10  in.,  a  high  force  factor, 
and  a  voice  coil  7/ie  in.  long  moving 
in  a  gap  with  a  3/s-in.  top  plate. 

The  curves  are  typical  of  commer- 
cial direct  radiator  speakers.  Ap- 
proximately half  the  distortion  at 
low  frequencies  is  due  to  the  non- 
linear force  displacement  curves  of 
the  diaphragms.  The  remainder  is 
due  to  the  non-linear  average-fhix- 
displacement  curves  of  the  driving 
mechanism. 

Since  these  curves  are  representa- 
tive of  commercial  practice  they  are 
of  interest  in  indicating  the  amount 
of  distortion  the  untrained  listener 
will  tolerate.  Speakers  designed  to 
give  less  distortion  for  a  given  cost 
will  have  less  loudness  efficiency,  and 
less  apparent  bass  response. 

Distortion  curves  of  horns  show 
distortion  of  the  same  order  in  the 
middle  frequency  range.  About  a 
third  of  an  octave  above  the  theoret- 
ical cutoff  frequency,  the  distortion 
begins  to  rise,  and  below  cutoff  usu- 
ally considerably  exceeds  the  distor- 
tion for  direct  radiator  speakers. 
This  distortion  largely  accounts  for 
the  horn's  sounding  as  if  it  radiated 
effectively  below  its  cutoff  frequency. 
For  a  more  complete  discussion  of 
testing  methods  see  the  references  in 
the  bibliography  marked  "tests." 

Another  measure  of  distortion  is 
the  intermodulation  of  two  signals 
of  non-harmonic  frequencies  produc- 
ing inharmonic  modulation  products. 
Because  the  annoyance  of  this  type 
of  distortion  is  extremely  high,  much 
smaller  amounts  may  be  tolerated 
than  with  non-linear  distortion. 
Though  the  technique  is  not  yet 
standardized,  a  useful  procedure  is 
to  fix  a  low-frequency  signal  at  a 
large  value  while  a  variable  high- 
frequency  scanning  signal  of  much 
smaller  value  is  applied  through  a 
linear  combining  circuit.  A  plot  of 
the  sound  pressure  due  to  the  scan- 
ning signal  and  that  due  to  the 
modulation  products  affords  a  means 
of  assessing  the  audibility  of  the 
modulation  products.  It  is  impor- 
tant that  this  be  done  as  a  continu- 
ous function  of  frequency  in  order 
not  to  miss  regions  of  large  distor- 
tion. This  is  shown  in  Fig.  10  for 
an  experimental  direct  radiator 


40   50  60      80  100        ISO  200       300  400  500 
Frequency  fa  Cycles  per  Second 


Fio.  9.    Steady-state  Non-linear  Distortion  in  Moving- 
coil  Direct  Radiator  Speaker 


BIBLIOGRAPHY 


13-21 


with  a  12-db  difference  between  power  available  for  low-  and  high-frequency 
signals. 

The  power  rating  of  a  loudspeaker  is  intended  to  provide  the  user  with  a  yardstick  by 
which  he  may  set  operating  conditions  to  insure  trouble-free  life  at  any  input  up  to  the 
maximum.  While  this  rating  may  be  limited  by  distortion,  thermal  overload,  or  mechan- 
ical overload,  the  latter  two  are  usually  controlling.  Thus  the  power  rating  usually  sets  a 
limit  above  which  mechanical  failure  may  be  expected  to  occur.  In  one  method  of  specify- 
ing this  power,  a  saturating  signal  of  a  variety  of  program  material  is  supplied  to  an  am- 
plifier of  adjustable  saturation  power,  used  as  the  source.  The  rating  power  is  that  below 
which  failure  will  not  occur  within,  say,  100  hours  for  at  least  90  per  cent  of  a  group  of 
speakers.  Thus  the  rating  is  essentially  in  terms  of  an  amplifier  whose  output  will  be 
safely  handled.  In  another  method  of  rating,  the  amplifier  has  a  much  greater  rating  power 
than  the  speaker  and  is  supplied  a 
synthetic  signal,  such  as  noise,  warble 
tone,  or  multitone.  These  have  rea- 
sonably constant  spectral  composition 
and  peak-to-rms  ratio,  and  the  power 
is  increased  until  the  100-hour  failure 
point  is  reached.  Though  neither 
method  is  yet  standardized,  it  is 
nevertheless  the  intent  of  both  to 
provide  a  realistic  and  useful  value. 

A  final  important  characteristic  of 
a  speaker  is  its  impedance.  Both 
modulus  and  angle  usually  exhibit 
considerable  variation  with  frequency, 
and  the  problem  is  to  connect  this 
load  to  a  resistive  source  of  essentially 
constant  impedance.  When  properly 
connected,  the  source  is  able  to  deliver 
to  the  load  the  maximum  program 
energy  consistent  with  distortion  re- 
quirements. In  practice  this  "con- 
nection" impedance  is  selected  by  the 
manufacturer  on  the  basis  of  his  listen- 
ing tests  and  experience.  It  may  be 
called  the  rating  impedance  of  the 
speaker,  and  it  indicates  the  tap  on 

the  amplifier  output  to  which  the  speaker  should  be  connected.  It  is  not  necessarily  the 
value  at  400  cps,  or  the  minimum  in  direct  radiator  speakers.  Besides  its  use  in  deter- 
mining speaker  connections,  the  rating  impedance  is  often  used  as  the  generator  resist- 
ance in  speaker  tests. 


1—20 


400       700 


2K  4K 

Frequency  .  CPS 


20K 


FIG.  10.    Modulation  Products  for  Experimental  12-in. 

Direct-radiator  Speaker  in    Infinite  Baffle.     Available 

power  input,  100  cps  fixed,  16  watts;  scanning  signal,  1 

watt. 


BIBLIOGRAPHY 

American  Standards  ^Association,  Standard  C16.4-1942.     (Tests.) 

Ballantine,  S.,  Proc.  I.R.E.,  Vol.  23,  618  (1935).    (Tests.) 

Barnes,  E.  J.,  W.B.,  VoL  7,  248,  301  (1930).    (Tests.) 

Barrow,  W.  L.,  J.A.S.A.,  Vol.  3,  562  (1932). 

Bostwick,  L.  GM  J.A.S.A.,  VoL  2,  242  (1930).     (High-frequency  horn  unit.)     B.S.T.J.,  VoL  8,  135 

(1929)      (Speaker  tests.) 

Von  BraunmuhL  H.  J.t  and  W.  Weber,  AJbw.  Zeits.,  VoL  2,  135  (5/37).     (Annoyance  of  distortion.) 
Chinn,  H.  A.,  and  P.  Eisenberg,  Proc.  I.R.E.,  VoL  33,  571  (9/45).    (Listener  preferences.) 
Cosers,  C.  R.,  W.E.,  VoL  6,  353  (1929).    (Hornless  speaker.) 

Davis,  A.  H.,  Modern  Acoustics.    (General.)  ,„«„,« 

Edelman  P  E    Condenser  Loud  Speakers  with  Flexible  Electrodes,  Proc,  I.R.E.,  VoL  19,  256  (1931). 
Fletcher,  H.,  Proc.  I.R*E.,  Vol.  30,  266  (6/40).    (Role  of  hearing,) 

Gerlach,  E.,  Phynk.  Zeit.,  VoL  25,  675  (1924).    (Ribbon  speaker.)  J0_,_       -        T  w 

Greaves,  V.  F.,  F.  W.  Kxanz,  and  W.  D.  Krozier,  The  Kyle  Condenser  Loud  Speaker,  Proc.  I.R.E., 

Hannie   W.,  Wi&senschafitickt  VerSffenttickungen  aus  dem  Siemens  Konzem,  VoL  10,  73  (1931),  abo 
VoL  11,  1  (1932).    (Hornless  and  condenser  speaker.)  , 

Hanna,  C.  R.,  Proc.  I.R.E.*  Vol.  13,  437  (1925).    (Balanced-armature  magnetic  speaker.)    J  AI.E.E 
VoL  47,  607  (1928).     (Horn  unit.)    Theory  of  the  Electrostatic  Loud  Speaker,  J.A.S.A.,  VoL  2,  143 

Hartmann,  C.  A.,  and  H.  Jacoby,  Elek.  Nach.  Tech.,  VoL  12,  163  (1935).    (Tests.) 
I.R.E.  Standards  on  Electroacousiics  (1938). 

Kellogg,  E.  W.,  J.A.S.A.,  VoL  2,  157  (1930)  (speaker  tests);  VoL  3,  94  (1931)  Cow-frequency  radia- 
tion) ;  VoL  4,  56  (1932). 

Kennelly,  A.  E.,  Electrical  Vibration  Instruments.    (Tests.) 
Knowles,  H.  S.t  Electronics,  VoL  4,  154  (1932);  VoL  6,  240  (1933).    (Tests.) 


13-22  ELECTROMECHANICAL-ACOUSTIC  DEVICES 

McLachlan,  N.  WM  Speakers,  J.A.S.A.,  VoL  5,  167  (1933)  (tests,  speakers,  and  bibliography);  W.  B.t 

VoL  9,  329  (1932),  Vol.  10,  204,  375  (1933)  (distortion.) 
Meyer,  E.,  Electroacoustics.    (Tests,  general.) 


N^poL  11,  548  (1929);  WissenschaMicke  VerQff^Uchungen  aus  dem 
Siemens-Konzern,  VoL  9,  226  (1930);  and  Siemens  Zeitscknft,  Vol.  10,  562  (1930).  (Speakers  and 

Oliver!  D.  A.,  IF.  B.,  Vol.  7,  653  (1930);  Vol.  10,  420  (1933) 

Olney,  B.f  Proc.  J.fl.k,  Vol.  19,  1113  (1931)  (speaker  tests);  /.A.S.A.,  Vol.  8,  104  (10/36)  (enclosures). 

Olson,  H.  F.,  Acoustical  Engineering.     (Construction,  tests,  general.)  MOOO  /-*  i     u 

Olson  H.  F.,  and  F.  Massa,  Applied  Acoustics  (speakers-tests);  J.A.S.A.,  Vol.  6,  250  (1935)  (telephone 
receivers)  ;  Proc.  I.R.E.,  VoL  21,  673  (1933)  (telephone  receivers.) 

Pedersen,  P.  0.,  J.A.S.A.,  Vol.  6,  227  (1935);  Vol.  7,  64  (1935).    (Subharmomcs  theory.; 

Rice,  C,  W,,  and  E.  W  Kellogg,  /.  AJ.E.E.,  Vol.  44,  982,  1015  (1925).     (Hornless  m.c.  speaker.) 

Schafer,  0.,  Hochf.  u.  Elek.,  VoL  44,  101  (1934).    (Speaker  test.) 

Sehaffstein,  G.,  Hochf.  u.  EUk.,  VoL  45,  204  (1935).    (Speaker  and  tests.) 

Schottky,  W.,  Physik  Zeit.,  VoL  25,  672  (1924).    (Ribbon  speaker.) 

Seabert,  J.  D.,  Proc.  I.R.E.,  VoL  22,  738  (1924).    (Speakers.)  . 

Stenzel,  H.,  Handbuchder  experimental  Physik,  VoL  17/2,  254  (1933)  (loudspeaker,  general)  jSZecirtscAe 
Nach.  Tech.,  Vol.  4,  239  (1927);  Vol.  6,  165  (1929);  VoL  7t  90  (1930);  also,  Zwt.  tech.  Physik,  VoL 
10,  569  (1929),  and  Ann.  der  Physik,  VoL  11,  947  (1930)  (directivity  of  radiators). 

Vogt,  H.,  Elec.  tech.  Zeit.,  VoL  52,  1402  (1931).    (Test  condenser  speaker.) 

Von  Sehmoller,  F.,  Telefunken  Zeit.,  VoL  15,  47  (1934).    (Subharmonics.) 

Wagner,  K.  W.,  Die  wissenschaftlichen  Grundlagen  des  Rundfunkempfangs.     (General  and  speakers.) 

Wegel,  R.  L.,  J.  AJ.E.E,,  Vol.  40,  791  (1921).     (Moving-coil  and  magnetie-armature  theory.) 

Wente,  E.  C.,  and  A.  L.  Thuras,  B.S.T.J.,  Vol.  7,  140  (1928)  (horn  unit);  Vol.  13,  259  (1934). 

Wheeler,  H.  A.,  and  V.  E.  Whitman,  Proc.  I.R.E.,  Vol.  23,  610  (1935).    (Test.) 

Williams,  S.  T.,  J.  Franklin  Inst.,  VoL  202,  413  (1926).    (Horn  speaker.) 

Wolff,  I.,  and  L.  Malter,  J.A.S.A.,  VoL  2,  201  (1930).    (Directivity  of  radiators.) 

Wood,  A.  B.,  Sound.     (GeneraL) 


MICROPHONES 

By  Hugh  S.  Knowles 

A  microphone  is  an  electroacoustic  transducer  actuated  by  energy  from  an  acoustical 
system,  and  delivering  energy  to  an  electrical  system,  the  spectral  composition  of  the  energy 
in  the  two  systems  being  substantially  equivalent.  They  may  be  classified  as  to  reversi- 
bility, type  of  generator,  acoustical  quantity  to  which  the  output  is  proportional,  and  type 
of  directional  characteristic. 

Reversible  microphones  are  those  that  are  also  capable  of  converting  electrical  into 
acoustical  energy,  and  they  include  moving-conductor  (ribbon  and  moving-ooil) ,  variable- 
capacitance  (condenser),  variable-reluctance  (moving-iron)  and  piezoelectric  (crystal) 
generator  types.  The  most  important  irreversible  microphone  is  the  carbon  (variable- 
resistance)  type, 

Reversible  microphones  may  be  treated  analytically  by  the  same  equations  used  for 
the  corresponding  loudspeakers,  provided  that  the  emf  in  the  electrical  mesh  is  set  equal 
to  zero  and  a  force  is  introduced  into  the  mechanical  mesh  (see  Loudspeakers,  p.  13-11). 

12.  FORCE  ON  THE  MICROPROS 

In  practically  all  microphones  the  electrical  output  is  proportional  to  the  net  force  on 
the  generating  element;  less  common  types  are  those  directly  sensitive  to  the  particle 
velocity  (hot-wire)  and  to  the  temperature  (pyroelectric)  of  the  sound  wave. 

PRESSURE  MICROPHONES.  In  pressure-responsive  microphones  the  active  sur- 
face is  moved  by  a  force  which  is  the  integral  of  the  pressure  over  it.  The  pressure  will 
deviate  from  that  in  the  free  wave  owing  to  the  effects  of  diffraction,  cavity  resonance, 
and  mechanical  impedance  of  the  active  surface.  The  last  effect  is  usually  small,  and  the 
others  may  be  minimized  or  in  some  instances  utilized  by  careful  attention  to  the  size 
and  shape  of  the  microphone.  The  microphone  response  may  be  stated  in  terms  of  the 
undisturbed  pressure  in  the  wave,  or  in  terms  of  tbe  pressure  at  the  active  surface;  these 
are  respectively  the  field  and  the  pressure  response. 

Diffraction  effects  are  important  when  the  maximum  dimension  of  the  microphone 
becomes  much  greater  than  a  tenth  wavelength.  For  frequencies  well  above  this  region 
the  pressure  will  be  doubled  (6  db)  over  the  undisturbed  value  unless  the  diaphragm  im- 
pedance is  very  low.  Since  diffraction  effects  are  also  functions  of  angle,  large  pressure- 
sensitive  microphones  may  be  expected  to  be  directional  at  the  higher  frequencies.  Screens 
are  sometimes  used  to  modify  the  diffraction  effects. 

When  the  active  surface  is  recessed,  the  shallow  cavity  will  exhibit  a  broad  resonance 
which  may  be  used  to  improve  the  high-frequency  response.  Though  "gains"  of  the  order 


MOVING-CONDUCTOR  MICROPHONES 


13-23 


Gridded  Cap 


Diaphragm 


of  5  db  may  be  attained,  the  sensitivity  to  angle  again  introduces  directivity.    If  the  force 
resulting  from  a  wave  of  normal  incidence  is  to  vary  less  than  2  db  from  that  at  grazing 
incidence,  then  the  active  surface  should  be  recessed  no  more  than  0.1  its  diameter.    At 
the  highest  frequency  to  be  reproduced  the 
diameter  should  be  less  than  a  half  wavelength. 
In  Fig.  1  is  shown  a  miniature  condenser  micro- 
phone which  approximates  these  conditions  up 
to  over  3  kc. 

PRESSURE-DIFFERENCE  MICRO- 
PHONES. When  two  points  in  space  are 
separated  by  a  small  fraction  of  a  wavelength 
the  difference  in  pressure  at  these  points  ap- 
proximates the  product  of  the  distance  they 
are  separated  and  the  pressure  gradient  in  the 
sound  field.  For  this  reason  pressure-difference 
microphones  are  sometimes  loosely  called  pres- 
sure-gradient microphones.  (See  also  Ribbon 
Microphones,  in  article  13.) 

When  both  sides  of  the  active  surface  are 
exposed  to  a  sound  wave,  the  net  force  on  it 
depends  on  the  difference  between  the  forces  on 
the  two  sides.  Let  the  acoustic  path  length 
from  one  side  to  the  other  be  Ij,.  Then  when  a 
wave  of  undisturbed  sound  pressure  amplitude, 
p,  is  incident  on  the  microphone  at  an  angle  8  FIG.  1.  Miniature  CoadeEtser  Microphone 
to  the  axis  of  symmetry,  the  magnitude  of  the 

net  force,  /,  is,  assuming  no  gradient  along  the  active  surface,  and  also  assuming  that  the 
pressure  difference  may  be  replaced  by  the  gradient, 


If  wZ&  is  sufficiently  small, 


Ct^fr  COS  & 


cos  B 


(1) 
(2) 


Thus,  up  to  frequencies  for  which  the  path  difference  h  is  an  appreciable  fraction  of  a 
wavelength,  the  net  force  is  proportional  to  that  path  difference  and  to  the  frequency. 
When  the  wave  front  is  spherical,  the  pressure  gradient  exceeds  its  value  for  a  plane  wave 
by  the  factor  [1  -f  (c/wr)2]^,  which  is  small  for  <ar  large.  When  the  wave  front  is  that 
due  to  a  piston,  the  axial  gradient  contains  terms  dependent  on  the  ratio  of  piston  diameter 
to  microphone  distance.  However,  when  this  ratio  is  less  than  0.5,  the  wave  front  is 
sufficiently  spherical  to  permit  the  spherical  wave  correction  factor  to  be  used. 

FLUID  IMPEDANCE.  Studies  of  reciprocity  laws  indicate  that  the  force  of  the 
wave  is  applied  through  a  "generator  impedance"  equal  to  that  into  which  the  active 
surface  works  when  radiating  the  same  wave  fronts  as  those  applied.  See  article  2  for 
results  for  some  simple  shapes.  In  general  this  added  impedance  is  predominantly  mass- 
like  up  to  the  high-frequency  region. 


13.  MOVING-CONDTICTOR  MICROPHONES 

In  this  class  electrical  energy  is  generated  by  the  motion  of  single  or  multiple  conductors 
in  a  magnetic  field,  as  exemplified  by  ribbon  and  moving-coil  types. 

RIBBON  MICROPHONES.  The  more  important  constructional  details  of  a  ribbon 
microphone  are  shown  in  Fig.  2.  The  moving  ribbon  element  is  both  diaphragm  and  con- 
ductor; it  may  be  0.312  in.  by  2  in.  by  0.0002  in.  and  is  corrugated  to  minimize  spurious 
modes  of  vibration.  As  a  low-frequency  approximation,  the  effect  of  the  fluid  impedance 
is  to  add  a  mass  of  the  order  of  magnitude  of  that  of  the  ribbon;  this  added  mass  decreases 
at  higher  frequencies.  The  mechanical  system  is  resonated  at  a  frequency  below  the  range 
to  be  reproduced,  resulting  in  a  mass-controlled  device. 

In  pressure-difference  responsive  ribbon  microphones,  both  sides  of  the  ribbon  are  ex- 
posed to  the  wave.  Thus  the  net  force  at  low  frequencies  is  roughly  proportional  to  the 
pressure  gradient  and  hence  to  the  velocity  in  a  plane  wave.  For  this  reason  the  term 
velocity  is  sometimes  applied  to  microphones  of  this  type.  However,  it  appears  desirable 
to  reserve  the  term  velocity  to  describe  devices  such  as  the  Rayleigh  disk  in  which  the 


13-24  ELECTROMECHANICAL-ACOUSTIC   DEVICES 

actuating  force  arises  more  directly  from  the  particle  velocity.  Up  to  fairly  high  fre- 
quencies both  the  force  and  impedance  (mass  controlled)  are  proportional  to  frequency, 
leading  to  a  constant  ribbon  velocity  and  hence  constant  generated  emf.  Control  of  the 
response  beyond  this  point  is  achieved  by  adjusting  the  size  and  disposition  of  the  pole 
pieces  and  other  elements  separating  the  two  faces  of  the  ribbon. 

If  one  side  of  a  ribbon  microphone  is  shielded  from  the  wave  by  a  damped  enclosure, 
the  device  becomes  pressure-responsive.  With  a  sound  pressure  independent  of  frequency 
the  output  is  independent  of  frequency  if  the  enclosure  adds  sufficient  resistance  to  make 

the  effective  impedance  of  the  diaphragm  substantially 
resistive.  Such  a  pressure-responsive  ribbon  microphone 
is  used  principally  with  a  pressure-difference  responsive 
type  in  a  directional  combination. 


FIG.  2.     Pressure-difference  Rib- 
bon Microphone. 


FIG.  3.     Moving-coil  Microphone 


MOVIN"G-COIL  MICROPHONE.  When  the  moving  conductor  is  a  moving  coil, 
there  results  a  form  similar  to  that  of  a  moving-coil  loudspeaker,  as  in  Fig.  3.  With  the 
net  force  in  the  mid  and  high  frequency  ranges  largely  due  to  that  on  the  outside  of  the 
diaphragm,  the  device  is  pressure-responsive  in  these  ranges.  In  order  to  obtain  uniform 
coil  velocity  and  hence  response  the  moving  system  is  designed  to  have  substantially 
uniform  impedance.  The  diaphragm  and  coil  assembly  is  resonated  in  the  low-middle 
frequency  range. 

Coupled  circuits  are  used  to  maintain  uniform  response  at  the  extreme  frequencies.  In 
Fig.  3  the  circuit  elements  leading  through  the  tube  t  to  the  rear  of  the  diaphragm  are  so 
chosen  as  to  achieve  an  increasing  force  at  low  frequencies  to  offset  the  increasing  stiffness 
reactance  of  the  diaphragm.  At  high  frequencies  the  stiffness  reactance  of  the  air  film 
under  the  dome  and  the  impedance  of  the  tuned  cavity  in  the  pole  tip  modify  the  impedance 
of  the  diaphragm  and  coil  assembly  so  as  to  maintain  approximately  uniform  velocity. 
Thus  it  is  possible  to  attain  a  fairly  uniform  response  from  40  to  10,000  cps. 

It  is  possible  to  combine  the  output  of  adjacent  pressure  and  pressure-difference  micro- 
phones in  such  a  manner  that  directional  patterns  are  obtained  which  are  members  of  the 
limaQon  family  of  which  the  cardioid  is  a  special  case.  Common  types  use  moving-coil 
pressure  and  ribbon  pressure-difference  microphones  connected  through  suitable  phase 
and  amplitude  controlling  networks. 


14.  CONDENSER  MICROPHONES 

Most  variable-capacitance  microphones  are  made  pressure-responsive,  with  a  mechan- 
ical system  tuned  above  the  desired  range.  A  frequency-independent  driving  force  acting 
upon  this  stiffness-controlled  system  results  in  a  displacement  independent  of  frequency. 
Since  the  output  emf  is  closely  proportional  to  the  displacement,  a  flat  pressure  response 
is  approached. 

To  the  mass  of  the  diaphragm  is  added  the  radiation  mass,  and  to  its  stiffness,  that  of 
the  air  film  between  diaphragm  and  rear  electrode.  See  Fig.  1.  The  diaphragm,  is  of  a 
high-tensile-strength  aluminum  or  stainless-steel  alloy  of  the  order  of  0.00075  in.  thick, 
and  is  tuned  by  stretching,  the  smaller  diaphragms  being  tunable  to  higher  frequencies. 
The  resistance  of  the  thin  interelectrode  air  film  materially  reduces  the  magnitude  of  the 
high-frequency  resonant  peak. 

The  small  cavity  and  the  high  mechanical  impedance  of  the  condenser  microphone 
make  it  particularly  suitable  for  calibration  by  reciprocity  methods.  Two  similar  micro- 
phones whose  comparative  response  is  known  are  placed  in  a  closed  cylindrical  volume  of 
which  each  forms  an  end;  the  output  of  one  is  noted  when  the  other  is  electrically  driven. 


CBYSTAL  MICROPHONES 


13-25 


As  with  all  reciprocity  methods,  its  success  depends  on  the  presence  of  a  definite  and 
calculable  acoustic  element,  which  here  is  the  compliance  of  the  common  volume,  corrected 
for  microphone  mechanical  impedance.  With  careful  control  of  acoustic  and  electrical 
parameters,  excellent  reproducibility  of  results  is  attained.  The  upper  frequency  limit 
may  be  extended  to  15,000  or  more  cps  by  the  use  of  hydrogen  hi  which  the  wavelength  is 
increased  by  a  factor  of  nearly  4. 

As  reciprocity  methods  yield  a  pressure  calibration,  a  correction  for  cavity  resonance 
and  diffraction  is  necessary.  When  the  condenser  microphone  is  used  as  a  free  field  meas- 
urement standard,  it  is  common  practice  to  utilise  these  factors  to  extend  the  frequency 
range,  with  electrical  equalizers  added  for  smooth  overall  response. 

15.  MAGNETIC-ARMATURE  MICROPHONES 

A  magnetic-armature  microphone  is  a  moving-iron  (variable-reluctance)  device  which 
finds  greatest  application  where  raggedness  and  high  output  are  the  main  desiderata.  An 
important  use  is  in  sound-power  telephones  (using  no  auxiliary  source  of  power,  such  as 
batteries),  which  place  a  premium  on  maximum  transmission  of  intelligence.  This  is 
achieved  by  resonating  the  moving  system  at  two  or  more  frequencies  in  the  1-3  kc  band 
in  which  maximum  articulation  is  obtained  for  a  given  amount  of  energy.  (For  force 
factor  and  force  equations  see  article  7,  Magnetic-armature  Speakers.) 


16.  CRYSTAL  MICROPHONES 

In  this  type  use  is  made  of  the  piezoelectric  properties  of  Rochelle  salt,  ammonium 
dihydrogen  phosphate,  or  more  rarely  tourmaline  and  quartz.  The  sensitivity  may  be 
expressed  in  terms  of  the  charge  released  for  a  given  displacement  of  the  driving  point; 
this  depends  but  slightly  on  temperature.  However,  the  dielectric  constant  of  Rochelle 
salt  and  hence  the  capacitance  and  emf  generated  changes  considerably  with  temperature. 
The  effect  of  this  capacitance  change  is  minimized  by  operating  the  crystal  in  a  high- 
impedance  circuit,  or  less  frequently  by  shunting  it  with  a  fixed  capacitance.  The  max- 
imum operating  temperature  range  is  40  to  +130  deg  fahr,  and  maximum  output  occurs 
at  about  72  deg  fahr. 

The  maximum  operating  temperature  range  of  ammonium  dihydrogen  phosphate  is 
—40  to  +185  deg  fahr.  While  the  emf  generated  is  greater  than  for  Rochelle  salt,  the 
smaller  dielectric  constant  results  in  such  small  capacitances  that  the  full  output  is  difficult 
to  realize.  The  high  resistances  necessitated  by  the  small  capacitances  also  give  rise  to 
electrical  noise  problems. 

Since  the  emf  developed  by  crystals  depends  on  the  displacement,  a  stiffness-controlled 
system  in  which  the  displacement  is  substantially  independent  of  frequency  is  used.  When 
the  microphone  consists  of  a  pah*  of  composite  plates 
arranged  as  in  Fig.  4  to  form  a  sound  cell,  the  resonant 
frequency  may  range  from  8  to  over  40  kc. 

When  a  single  composite  plate  is  clamped  at  three 
corners  and  the  fourth  is  driven  by  a  small  diaphragm, 
the  output  voltage  increases  some  15  db  over  the  sound 
cell  construction,  but  the  resonant  frequency  is  then  re- 
duced to  a  few  kilocycles.  The  main  use  of  this  type  is 
for  speech,  in  which  case  the  output  is  allowed  to  rise 


Composite  Plates 


Sealing  Membrane 


Mounts,  Separating 
and  Damping  Slabs 

FIG.  4.     Crystal  Microphone  ("Sound  Cell"  Type) 


Plate 


FIG.  5.  Unidirectional  Diapbragm- 
driven  Crystal  Microphone 


substantially  at  the  fundamental  resonant  frequency  of  3  or  4  kc.  Auxiliary  acoustic 
meshes  are  used  to  obtain  smooth  response  with  reduced  sensitivity  in  broad-band  types 
used  for  music. 


13-26  ELECTROMECHANICAL-ACOUSTIC  DEVICES 

A  single  transducer  element  crystal  unidirectional  microphone  is  shown  in  section  in 
Fig.  5.  When  the  slot  is  closed  the  microphone  operates  as  a  pressure  type.  (See  Direc- 
tional Characteristics,  below.) 

17.  CARBON  MICROPHONES 

The  carbon  microphone,  the  most  important  irreversible  microphone,  is  a  ^variable- 
resistance  type  involving  loose  carbon  contacts.  The  sound  wave  actuates  a  diaphragm 
which  exerts  a  varying  pressure  on  a  large  number  of  fine  carbon  granules,  thus  modulating 
a  bias  current  obtained  from  a  d-c  source.  Usually  the  resistance  and  output  voltage  are 
approximately  proportional  to  the  displacement,  and  so  a  stiffness-controlled  system  is 
employed.  Because  of  the  difficulty  of  obtaining  uniform  response,  the  present^  uses  of 
carbon  microphones  are  restricted  to  applications  in  which  high  output  and  "crisp  "speech 
quality  are  paramount,  as  in  military  equipment.  Another  factor  militating  against  its 
use  for  high-fidelity  applications  is  the  low  signal-to-noise  ratio,  due  to  the  random  varia- 
tion in  contact  resistance  always  present. 

The  small  allowable  size  of  the  carbon  button  has  been  used  to  advantage  in  a  close 
talking  pressure-difference  microphone  with  marked  noise-reducing  properties.  Designed 
to  be  worn  directly  in  front  of  the  mouth,  the  microphone  is  immersed  in  the  large  pressure 
gradient  field  of  the  talker,  while  the  ambient  noise  has  a  much  smaller  gradient.  Thus 
the  signal-to-ambient-noise  ratio  is  improved,  permitting  intelligible  communications  from 
such  noisy  locations  as  planes,  tanks,  and  engine  rooms. 

18.  DIRECTIONAL  CHARACTERISTICS 

It  is  often  important  to  collect  sound  arriving  from  a  desired  region  to  the  exclusion 
of  randomly  incident  sound,  as  in  sound-reinforcing  systems.  The  ability  of  the  micro- 
phone to  accomplish  this  is  determined  by  the  dependence  on,  angle  and  frequency  of  the 
field  response.  There  are  many  measures  of  directionality,  such  as  the  ratio  of  output  for 
sound  from  the  desired  direction  to  the  output  for  sound  of  random  incidence  of  the  same 
total  power,  the  ratio  of  the  output  for  sound  of  random  incidence  in  the  front  hemisphere 
to  that  of  random  incidence  in  the  rear  hemisphere,  or  the  ratio  of  the  outputs  at  the  angles 
of  maximum  and  minimum  response. 

Most  commonly  used  microphones  may  be  classified  as  non-directional,  bidirectional, 
and  unidirectional.  Since  pressure  is  a  scalar  quantity,  an  ideal  pressure  microphone  is 
non-directional.  In  practice  complete  freedom  from  directional  effects  is  achieved  only 
when  the  maximum  transverse  dimension  is  of  the  order  of  an  eighth  wavelength  or  less. 
Since  most  microphones  are  operated  at  frequencies  up  to  4000  or  more  cps,  this  require- 
ment necessitates  a  microphone  a  centimeter  or  less  in  diameter.  To  provide  adequate 
sensitivity  microphones  are  made  larger  than  the  non-directional  requirements  dictate 
and  commonly  have  a  diameter  of  2  to  5  cm.  In  pressure-actuated  microphones  this 
results  in  a  non-directional  microphone  at  low  frequencies  and  a  unidirectional  one  at  high 
frequencies.  Screens  are  sometimes  mounted  near  the  diaphragm  to  alter  the  sound  field 
and  make  the  microphone  less  directional. 

The  most  common  bidirectional  microphone  is  the  cosine  pressure-difference  or  "gra- 
dient" type  exemplified  by  the  ribbon  microphone  discussed  above. 

Most  unidirectional  microphones  have  a  directional  characteristic,  which,  if  taken  in  a 
plane  through  the  principal  axis,  is  given  by  eu  —  &n  +  e&  cos  0.  If  the  microphone  has 
separate  non-directional  (pressure)  and  bidirectional  transducer  elements,  en  and  ej,  are 
their  respective  voltages  and  BU  is  their  combined  voltage.  One  type  of  unidirectional 
microphone  is  a  pressure-difference  type  in  which  a  single  transducing  element  is  used 
with  a  network  to  shift  the  phase  of  one  of  the  pressures  to  alter  the  directional  charac- 
teristic. In  this  type  &n  corresponds  to  the  voltage  with  no  force  contribution  from  the 
rear  network  and  et,  to  the  voltage  when  the  rear  and  front  impedances  are  equal  and  the 
microphone  is  bidirectional.  By  adjusting  the  impedance  of  the  rear  network  the  relative 
values  of  en  and  ei  may  be  altered  to  give  various  directional  characteristics.  Single  trans- 
ducer microphones  of  this  type  are  commonly  made  with  crystal  or  moving-coil  generators. 

19.  PERFORMANCE  AND  TESTS 

The  most  important  performance  criteria  are  the  field  response  (see  Fig.  6),  directivity, 
impedance,  and  inherent  noise.  Less  important  for  general  applications  are  the  "dynamic" 
range  (the  range  from  minimum  pressure,  limited  by  electrical  noise,  to  maximum  pressure 


BIBLIOGRAPHY 


13-27 


limited  by  non-linearity  or  by  structural  strength),  the  ratio  of  the  responses  for  near  and 
distance  sources,  termed  the  proximity  index,  and  non-linear  distortion. 

OBJECTIVE  TESTS.  The  field  response  of  a  microphone  is  a  measure  of  the  elec- 
trical output,  for  a  specified  frequency,  when  immersed  in  a  plane  progressive  ware.  When 
the  effect  of  the  impedance  is  considered,  an  expression  of  the  form  20  log  E/p  —  10  log  R 
results,  in  which  E  is  the  open-circuit  emf,  in  volts,  generated  by  the  microphone;  p  is  the 
undisturbed  sound  pressure,  in  dynes/ cm2;  and  R  is  the  stated  impedance  of  the  micro- 
phone. This  relation  takes  into  account  the  effect  of  impedance  in  permitting  the  use  of 
step-up  transformers.  For  crystal  and  condenser  microphones,  R  is  sometimes  assigned  a 
value  corresponding  to  the  maximum  stated  value  of  transformer  secondary  impedance 
used  with  low-impedance  microphones,  usually  from  30,000  to  100,000  ohms.  This  value 
permits  a  fairer  compari- 
son of  all  types  in  terms 
of  the  amount  of  amplifi- 
cation necessary,  referred 
to  an  input  grid. 

The  directivity  is  usu- 
ally determined  from  field- 
response  curves  taken  at 
various  angles  of  inci- 
dence, or  from  the  field 
response  as  a  function  of 
angle  with  the  spectral 
compositiQn  of  the  test 
signal  held  constant. 
Single-frequency,  narrow- 
band  warbled  or  fre- 
quency-modulated and 
noise  test  signals  are  used. 
Unidirectional  micro- 


-F5 


--5 


40    60     UDO 


10000  2OOOO 


200       4Q060O  1000     2QOQ     40OO 
Frequency  In  Cyctes  per  Second 

PIG.  6.     Axial  Response-freqiaency  Curves  of  Three  Microphones 


phones  should  show 
greater  than  10-db  front- 
to-rear  discrimination  (15  db  is  a  common  value),  while  in  the  bidirectional  (pressure- 
difference)  type  the  front-to-side  discrimination  usually  exceeds  20  db  over  a  large  fre- 
quency range. 

To  minimize  frequency  discrimination  and  reduce  electrostatic  and  electromagnetic 
induction  in  long  microphone  lines,  broadcast-type  microphones  have  impedances  ranging 
from  35  to  250  ohms,  with  a  trend  toward  150  ohms.  These  microphones  are  usually 
operated  into  impedances  ten  or  more  times  that  of  the  microphone's.  This  results  in 
about  a  3-db  improvement  in  the  signal-to-inherent-electrical-noise  ratio  over  that  ob- 
tained when  the  load  resistance  equals  the  modulus  of  the  microphone  impedance. 

The  present  technique  of  primary  calibration  is  by  means  of  the  reciprocity  technique 
mentioned  under  Condenser  Microphones,  article  14.  This  yields  a  pressure  calibration, 
from  which  the  field  response  may  be  derived  by  calculation  or  measurement  (with  scaled 
models)  of  diffraction  and  cavity  resonance  effects.  A  microphone  thus  calibrated  may 
be  used  as  a  standard  from  which  the  response  of  other  microphones  may  be  obtained  by 
comparison,  in  absolute  terms.  The  source  may  be  a  sufficiently  distant  loudspeaker  with 
very  smooth  response;  or  for  special  purposes  an  artificial  voice  is  used  to  obtain  field  shape 
and  diffraction  effects  approximating  those  of  a  person  speaking.  Although  it  is  possible 
to  obtain  free-field  reciprocity  calibrations,  much  work  remains  to  be  done  before  tbe 
precision  and  stability  of  the  pressure  reciprocity  method  can  be  attained. 

SUBJECTIVE  (AURAL)  TESTS.  As  with  loudspeakers,  the  final  acceptability  of  a 
microphone  depends  on  subjective  tests.  A  live  or  artificial  voice  may  be  used  as  a  source 
for  two  microphones  being  compared,  the  outputs  being  alternately  connected  to  an  addi- 
tion system.  Such  qualities  as  naturalness,  smoothness,  presence,  articralaton,  ®&  objexv 
tionability  of  intermodulation  distortion,  loudness  efficiency,  and  transient  distortion  may 
best  be  compared  by  aural  tests  with  a  trained  jury. 


BIBLIOGRAPHY 

Am.  Stds.  Assoc.  Standard  Z24.4-1938.     (Calibration.) 

Am  Stds  Assoc.  Standard  Z24.1-1942.    (Terminology.) 

Baerwald,  H.,  J.A.S.A.,  Vol.  12,  131  (1940).    (N°*e.) 

Ballantine,  S.,  J.A.S.A.,  Vol.  3,  319  (1932).    (Calibration.)      . 

Bauer   B.  B.,  J.A.S.A.,  VoL  13,  41  (1941).    (Unidirectional  microphone.) 


13-28  ELECTROMECHANICAL-ACOUSTIC  DEVICES 

Braunmuhl,  H.  J.  von,  and  W.  Weber,  Hochfrequenztechnik  u.  Elektroakustik,  Vol.  46,  187  (1935). 

(Directional  pressure-difference  condenser  microphone.)  . 

Cook,  R.  K.,  J.  Research,  National  Bur.  Stds.,  Vol.  25,  489  (1940),    (Reciprocity.) 
Crandall,  I.  B.,  Phys.  Rev.,  Vol.  11,  449  (1918).     (Condenser  microphone.)  . 

Efflthora,  H,  E.,  and  A.  M.  Wiggins,  PTOC.  I.R.E.,  Vol.  34,  84  (1946).    (Gradient  microphones.) 
Foldy,  L.  L.,  and  H.  Primakoff,  J.A.S.A.,  Vol.  17,  109  (1945).    (Reciprocity.) 
Gerlach,  E,,  and  W.  Schottky,  Physik  Zeits.,  Vol.  25,  276  (1923).    (Ribbon  microphone.) 
Glover,  R.  P.,  J.A.S.A.,  Vol.  11,  296  (1940).     (Cardioid  unidirectional  microphones.) 
Harrison  H  fc  ,  and  P  B.  Flanders,  B.S.T./.,  Vol.  11,  451  (1932).    (Miniature  condenser  microphone.) 
Marshall,  R.  N.,  and  W.  R.  Harry,  J.  S.M.P.E.,  33,  54  (1939).    (Cardioid  microphone.) 
Massa,  P.,  J.A.S.A.,  Vol.  17,  29  (1945).    (ADP  microphone.) 
Olson,  H.  F.,  Elements  of  Acoustical  Engineering.    (General.) 
Olson,  H.  F.,  R.C.A.  Rev.,  Vol.  6,  36  (1941).    (Reciprocity.) 
Sawyer,  C.  B.,  Proc.  I.R.E.,  VoL  19,  2020  (1931).    (Crystal  microphones.) 
Schottky,  W.,  Zeits.  f.  Physik,  Vol.  36,  689  (1926).     (Reciprocity.)     _  . 

Wente,  E.  C.,  and  A.  L.  Thuras,  J.A.S.A.,  Vol.  3,  44  (1931).    (Moving-coil  microphones.) 


MAGNETIC  RECORDING  AND  REPRODUCING 
OF  SOUND 

By  L.  Vieth  and  H.  A.  Henning 

The  earliest  record  of  magnetic  recording  is  credited  to  Poulsen,  who  in  1900  described 
his  "Telegraphone."  Since  that  time,  though  much  has  been  learned  and  vastly  improved 
results  have  been  obtained,  extensive  use  of  the  method  as  a  recording  process  of  great 
potentialities  was  not  achieved  until  the  war  years  of  1941-1945. 

The  magnetic  recording  method  possesses  certain  unique  advantages.  The  mechanism 
may  be  simple  to  operate  and  rugged.  The  record  can  be  played  immediately  after  record- 
lag  and  can  be  replayed  practically  any  number  of  times.  The  recording  may  be  erased 
and  the  medium  reused  as  often  as  desired. 

Magnetic  recording  involves  three  fundamental  operations  or  processes:  (1)  erasing; 
(2)  recording;  (3)  reproduction  or  playback.  Erasing  is  the  process  by  which  the  magnetic 
recording  medium  is  either  neutralized  or  saturated  to  obliterate  any  signal  previously 
recorded.  In  the  early  phases  of  development  saturation  by  a  strong  d-c  field  was  used. 
More  recently  neutralization  of  the  medium  by  a  high-frequency  a-c  field  has  been  required 
in  conjunction  with  modern  methods  of  recording  with  an  a-c  bias.  Recording  is  accom- 
plished by  applying,  through  the  recording  head,  the  signal  to  be  recorded  superimposed 
on  a  biasing  current.  The  bias  current  may  be  either  direct  or  high-frequency  alternating 
current.  Considerations  governing  the  choice  of  biasing  and  erasing  methods  are  discussed 
in  article  21.  The  bias  current  is  necessary  to  obtain  a  faithful  recording  of  the  impressed 
signals.  The  medium,  as  it  passes  under  the  influence  of  the  recording  head,  is  magnetized 
in  proportion  to  the  variations  of  the  signal  current,  and  this  magnetization  remains  until 
erased.  Reproduction  or  playback  is  accomplished  by  passing  the  recorded  medium  over 
a  magnetically  sensitive  head,  usually  at  the  same  velocity  as  in  recording.  The  voltages 
induced  in  the  head  are  then  amplified  and  equalized  to  obtain  the  desired  frequency  char- 
acteristic. All  these  processes  may  be  accomplished  with  the  same  head  at  some  sacrifice 
in  performance. 

20.  ERASING,  RECORDING,  AND  REPRODUCING  ARRANGEMENTS 

Physical  arrangements  by  which  the  magnetic  forces  are  applied  to  a  magnetic  medium 
(usually  in  the  form  of  a  tape  or  wire)  have  taken  several  forms.  They  may  be  divided 
into  three  broad  classifications:  (1)  perpendicular  magnetization,  in  which  the  direction 
of  magnetization  is  normal  to  the  surface  of  the  medium;  (2)  modifications  of  (1),  which 
alter  perpendicularity  somewhat;  and  (3)  longitudinal  magnetization  in  which  the  direction 
of  magnetization  is  parallel  with  the  direction  of  motion  of  the  medium.  Transverse  mag- 
netization in  which  the  direction  of  magnetization  is  parallel  to  the  surface  of  the  medium 
and  normal  to  the  direction  of  motion  is  a  fourth  possible  classification  which  will  not 
be  considered  here.  Figure  1  illustrates  schematically  the  three  arrangements.  Each 
consists  essentially  of  cores  of  high-permeability  material  surrounded  by  one  or  more 
coils.  Figure  l(a)  shows  the  perpendicular  application  of  magnetic  force  to  a  medium  in 
the  form  of  tape,  moving  between  the  poles,  each  consisting  of  one  lamination  in  exact 
alignment.  Each  pole  is  surrounded  by  a  coil.  A  concentration  of  recording  flux  is  obtained 
by  the  small  thickness  of  lamination  in  the  direction  of  tape  travel. 

Figure  1  (&)  illustrates  a  more  efficient  and  practical  modification  of  (a] .  Much  thicker 
cores  are  used,  which  are  in  contact  with  opposite  surfaces  of  the  tape.  Flux  concentration 
is  accomplished  by  offsetting  the  poles  in  the  direction  of  travel  by  an  amount  almost 


ERASING,  RECORDING,  REPRODUCING  PROCESSES      13-29 

equal  to  their  thickness.  A  coil  surrounds  the  advance  pole.  The  tape  is  therefore  mag- 
netized in  a  preponderantly  perpendicular  manner  with  a  longitudinal  component.  Induc- 
tion at  short  wavelengths  is  most  pronounced  on  the  side  of  the  tape  in  contact  with  the 
coil-bearing  pole.  Omission  of  the  coil  on  the  receding  or  following  pole  avoids  a  secondary 
concentration  of  recording  flu*  as  the  tape  leaves  the  influence  of  the  recording  head,  thus 
avoiding  modulation  of  the  already  recorded  signal.  The  schemes  illustrated  in  l<a)  and 
1W  require  that  intimate  contact  must  be  maintained  by  pressure  against  both  surfaces 
of  the  tape.  Joints  must  therefore  be  carefully  made,  and  the  two  surfaces  must  be  uni- 
formly parallel  to  insure  good  contact  at  all  times.  Furthermore,  these  two  schemes  are 
of  limited  use  on  wire  and  are  even  less  useful  on  non-magnetic  tapes  coated  on  one  side 
with  magnetic  materials.  Figure  l(c)  shows  an  arrangement  which  applies  magnetising 
force  longitudinally.  Contact  is  made  on  only  one  side,  and  the  principle  is  therefore 
conveniently  applicable  to  all  forms  of  recording  media  A  ring  of  permeable  material 


FIG.  1.     Typical  Magnetic  Recording  and  Reproducing  Arrangements 

wound  with  a  coil  is  provided  with  a  small  air  gap  at  the  point  where  it  touches  the  record- 
ing medium.  A  portion  of  the  flux  in  the  air  gap  passes  through  the  medium.  The  result- 
ing magnetization  is  substantially  longitudinal.  The  concentration  of  flux  is  accomplished 
by  the  use  of  very  small  gaps  ranging  from  0.003  in.  to  less  than  0.0003  in.,  depending  on 
the  medium. 

Although  all  the  above  arrangements  may  be  used  for  erasing,  recording,  and  reproduc- 
ing, it  is  expedient  to  alter  their  characteristics  in  the  interests  of  one  process  or  another. 
Separate  heads  are  frequently  provided  for  each  process.  In  each  case  it  is  important 
that  the  magnetic  action  of  the  head  in  recording  and  reproducing  be  concentrated  along  a 
line  perpendicular  to  the  direction  of  motion.  The  greatest  concentration  of  active  flux 
lines  is  obtained  for  a  given  pole-piece  arrangement  when  the  tip  reluctance  of  the  record- 
ing or  reproducing  pole  piece  is  low.  Poles  or  cores  are  usually  laminated  to  reduce  the 
frequency  discrirninating  effects  of  eddy  currents. 


21.  ERASING,  RECORDING,  AND  REPRODUCING  PROCESSES 

RECORDING  SPEED.  The  various  systems  used  for  driving  the  magnetic  media 
will  not  be  discussed  here.  There  are,  at  present,  no  standardized  recording  speeds,  aad 
in  most  applications  the  medium  is  moved  past  the  recording  and  reproducing  heads  at 
the  slowest  speed  that  will  insure  the  desired  high-frequency  response.  In  practice  these 
speeds  range  from  a  few  inches  to  several  feet  per  second.  In  order  to  obtain  a  freqtkeacy 
range  of  10,000  cycles  it  is  modern  practice  to  employ  recording  speeds  between  1  lfe  and 
5  ft  per  sec,  depending  on  the  properties  of  the  recording  medium  and  the  methods  of 
recording. 

ERASING.  The  process  used  to  obliterate  a  previously  recorded  signal  on  a  magnetic 
medium  is  called  erasing.  In  most  applications  the  erasing  operation  is  performed  during 
the  recording  process  by  a  head  located  somewhat  in  advance  of  the  recording  head.  There 
are,  however,  instances  in  which  it  is  more  convenient  to  perform  the  erasure  as  an  inde- 
pendent operation. 

A  previous  recording  may  be  removed  either  by  saturating  the  medium  with  a  strong 
d-c  field  or  by  neutralizing  the  medium  with  an  a-c  field  of  diminishing  intensity.  In  the 


13-30  ELECTROMECHANICAL-ACOUSTIC  DEVICES 


first  process  every  portion  of  the  medium  entering  the  erasing  head  is  carried  to  saturation 
or  a  degree  of  magnetization  exceeding  the  strongest  recorded  signal.  The  erasing  head 
may  consist  of  a  permanent  magnet  or  an  electromagnet  supplied  by  direct  current.  In 
certain  designs  the  configuration  of  leakage  flux  lines  around  the  head  can  cause  a  second 
and  reversed  field  to  act  on  the  medium  as  it  leaves  the  head.  Under  such  conditions  the 
initial  saturation  is  followed  by  a  partial  demagnetizing  operation,  and  the  ultimate  state 
of  the  medium  may  be  considerably  less  than  fully  saturated.  As  the  optimum  d-c  record- 
ing bias  is  determined  by  the  magnetic  state  of  the  erased  medium  it  is  apparent  that  the 
design  of  the  erasing  head  influences  the  constants  of  the  recording  process. 

It  is  now  common  practice  in  high-quality  magnetic  recording  to  erase  by  a  neutralizing 
or  demagnetizing  action.  This  is  accomplished  by  subjecting  each  element  of  the  medium 
to  a  cyclicly  varying  magnetic  field  whose  maximum  intensity  between  the  poles  of  the 
head  produces  saturation.  As  the  element  moves  away  from  the  head  it  is  subjected  to  a 
continuously  diminishing  cyclic  magnetization  which  leaves  the  medium  in  a  neutral  state. 
A  magnetizing  force  approximately  three  times  the  coercive  force  of  the  medium  has  been 
found  to  be  satisfactory.  A  field  intensity  of  1500  oersteds  may  be  required  to  demagnetize 
some  of  the  modern  high  coercive  force  materials. 

Some  use  has  been  made  of  an  air-core  coil  for  erasure  of  the  lower-coercive-force 
materials.  The  medium  is  drawn  through  the  center  of  the  coil,  and  a  comparatively 
low-frequency  erasing  current  is  required.  If  erasing  is  accomplished  with  the  recording 

head  a  higher-frequency 


RECORDfMG  FIELD,  H 


PIG.  2.    Magnetic  History  of  an  Element  of  Recording  Medium.    Sat- 
uration  erase,  d-c  Bias. 


current  is  used  to  provide 
the  necessary  number  of 
reversals  in  the  short 
time  the  medium  is  within 
the  influence  of  the  head. 

In  certain  instances  a 
combination  of  d-c  satu- 
ration followed  by  partial 
neutralization  has  been 
employed.  In  such  cir- 
cumstances the  recorded 
signal  is  successfully 
erased  but  there  remains 
a  residual  d-c  component 
on  the  medium  which  is 
responsible  for  somewhat 
increased  distortion. 

RECORDING.  The 
process  of  recording  con- 
sists of  impressing  on  the 
moving  magnetic  me- 
dium a  varying  induction 
which  is  directly  propor- 
^on-al  to  the  instantane- 
ous  value  of  the  recording 
head  current.  The 


. 

cording  head  is  usually  associated  with  an  amplifier  so  designed  that  the  recording  cur- 
rents are  independent  of  the  impedance  characteristic  of  the  head.  In  most  applications 
recording  currents  are  small  and  very  little  power  is  required.  In  certain  applications 
electronic  amplification  is  not  used  and  the  recording  current  is  obtained  directly  from  a 
carbon  microphone. 

The  linear  relationship  between  recording  current  and  the  resulting  magnetization  on  the 
medium  is  obtained  by  the  use  of  a  superimposed  biasing  field,  usually  by  superposing  a 
biasing  current  on  the  recording  current  fed  to  the  head.  Without  the  bias  the  reproduced 
signals  are  weak  and  very  distorted.  The  biasing  field  is  adjusted  to  a  magnitude  that 
minimizes  the  distortion  without  introducing  unnecessary  background  noise.  The  proper 
value  is  usually  slightly  less  than  that  which  will  yield  the  strongest  reproduced  signal. 
Two  biasing  methods  are  available.  It  is  now  customary  to  use  a  high-frequency  (super- 
sonic) biasing  current,  and  the  greatest  dynamic  recording  range  may  be  obtained  by  this 
method.  In  earlier  applications  a  d-c  biasing  current  was  used;  it  is  advantageous  where 
extreme  compactness  and  simplicity  of  equipment  are  essential. 

The  magnetic  processes  involved  during  recording  may  be  illustrated  by  hysteresis  loops 
such  as  those  of  Figs.  2,  3,  and  4.  These  curves  represent  Bi  (or  B-H}  vs.  H,  or  intrinsic 
induction  vs,  magnetizing  force.  They  thus  represent  the  fundamental  characteristic  of 


ERASING,   RECORDING,   REPRODUCING  PROCESSES      13-31 


the  recording  medium.  Each  element  of  the  medium,  as  it  passes  the  recording  point,  is 
subjected  to  a  magnetizing  field  which  is  proportional  to  the  algebraic  sum  of  the  instan- 
taneous recording  current  and  the  biasing  current.  The  element  is  then  withdrawn  from 
the  field.  For  long  uni- 
formly magnetized  sec- 
tions the  resultant  state 
of  each  element  of  the 
section  would  be  repre- 
sented by  a  point  on  the 
Bi  axis.  The  magnetic 
state  of  an  element  of  a 
short  magnet  (because  of 
self-demagnetization)  is 
represented  by  a  point  on 
an  appropriate  demag- 
netization coefficient  line. 
The  demagnetization  co- 
efficient line  OA  of  Figs. 
2,  3,  and  4  may  be  called 
the  open  circuit  line,  and 
it  defines  the  state  of  the 
element  when  removed 
from  the  head.  When- 
ever the  element  under 
consideration  is  within 
the  playback  head,  the 
self-demagnetizing  field 
is  partially  removed  and 
the  magnetic  state  of  an 
element  moves  along  a 


FIG.  3.    Magnetic  History  of  aa  Element  of  Recording  Medium.    Neu- 
tralization erase,  d-c  bias. 


reversible  minor  hysteresis  curve  which  may  be  represented  by  a  straight  line  such  as  Dd 
in  Fig.  2  to  a  steeper  demagnetization  coefficient  line  OB,  The  slope  of  line  OB,  which 
may  be  called  the  closed  circuit  line,  is  determined  primarily  by  the  reluctance  present  at 
the  contact  point  of  medium  and  head  and  thus  is  substantially  independent  of  the  recorded 

wavelength.  In  longitudinal 
recording  the  slope  of  the 
open-circuit  line  OA  is  to 
some  extent  a  function  of  the 
recorded  wavelength,  and 
thus  the  self-demagnetization 
is  a  function  of  frequency. 
The  effect  on  the  frequency 
response  of  this  action  may 
be  at  least  partially  offset*  by 
the  use  of  high-coercive-force 
materials. 

Figure  2  illustrates  the 
magnetic  process  of  recording 
with  a  d-c  bias  on  a  medium 
that  has  previously  been  sat- 
uration erased.  During  the 
erasing  process  the  saturating 
field  brings  the  magnetic  state 
of  each  element  of  tfee  medium 

\  to  a  point  C  on  the  hysteresis 

\  curve.      Assuming   that    the 

element  leaves  the  erasing 
field  without  being  subjected 
to  a  reverse  magnetization, 
the  magnetic  state  of  the  ele- 
ment moves  along  the  hys- 
teresis curve  CC'D  to  point 


PIG.  4.    Magnetic  History  of  an  Element  of  Recording  Medium* 
Neutralization  erase,  a-c  bias. 


D  on  the  open-circuit  line  OA.  Subsequently,  when  in  the  recording  head,  the  state  of 
the  element  is  represented  by  point  d  on  closed-circuit  line  OB,  and  as  the  minor 
hysteresis  curve  Dd  traversed  during  this  process  is  substantially  reversible  any  applied 


13-32  ELECTROMECHANICAL-ACOUSTIC  DEVICES 

recording  field  operating  within  this  region  leaves  no  change  in  impression  on  the 
medium.  If  after  saturation  a  d-c  bias  is  applied,  the  state  of  the  element  is  brought  to 
point  E  along  the  major  loop,  and  this  point  is  chosen  to  represent  the  midpoint  of  the 
available  operating  range.  Variations  in  the  recording  field  will  then  produce  correspond- 
ing changes  in  magnetization.  The  combined  action  of  the  biasing  field  and  the  signal 
field  then  operates  over  a  substantially  straight  section  DEF  of  the  hysteresis  curve.  When 
the  element  emerges  from  the  recording  head,  its  magnetic  state  returns  along  the  minor 
loop  to  the  open-circuit  line  OA  and  subsequently,  when  in  the  reproducing  head,  to  the 
closed-circuit  Hue  OB  along  the  same  minor  loop.  Depending  on  the  signal  strength,  the 
path  of  this  action  is  along  nearly  straight,  parallel,  and  reversible  minor  hysteresis  curves 
Ijing  between  Dd  and  FO.  Thus  the  flux  induced  during  reproduction,  for  the  state  of 
magnetization  corresponding  to  a  point  on  the  closed-circuit  line  OB  lying  between  0  and  d, 
is  practically  linearly  related  to  the  applied  recording  field. 

In  Fig.  2,  the  point  F  represents  one  of  the  limits  of  the  recording  field  which  may  be 
applied  to  the  medium  without  serious  distortion.  This  is  because  minor  hysteresis  curves 
have  greater  curvature  after  crossing  the  Bi  axis  and  a  minor  hysteresis  curve  such  as  Gh 
leaves  the  magnetic  state  of  the  element  at  gf  instead  of  the  required  point  g. 

The  operating  range  during  recording  at  short  wavelengths  is  also  restricted  wherever 
the  slope  of  the  open-circuit  line  OA  is  a  function  of  the  wavelength.  This  is  illustrated 
in  Fig.  2  by  line  OA',  which  might  be  the  slope  of  the  open-circuit  line  in  a  medium  longi- 
tudinally recorded  at  a  short  wavelength.  The  magnetic  state  of  a  saturated  element  is 
then  represented  by  the  point  Df.  The  recording  field  is  thus  restricted  to  operation  over 
the  region  between  D'  and  F. 

Figure  3  illustrates  the  magnetic  process  involved  when  a  d-c  bias  is  used  with  a  com- 
pletely demagnetized  medium.  The  recording  bias  alone  serves  to  carry  the  magnetic 
state  of  the  element  along  the  normal  magnetization  curve  ODEF  to  point  E.  The  record- 
ing signal  operates  about  Et  which  is  chosen  as  the  midpoint  of  the  linear  part  of  the  mag- 
netization curve.  Upon  leaving  the  recording  head  the  magnetization  of  the  element 
drops  to  line  OA  along  one  of  substantially  parallel  hysteresis  curves  similar  to  and  lying 
between  Dd'  and  F/'.  The  process  is  thereafter  similar  to  that  described  for  saturation 
erase.  If,  at  shorter  wavelengths,  the  slope  of  the  open-circuit  demagnetization  coefficient 
line  OA  is  reduced  to  OA',  points  d  and  /  on  the  closed-circuit  demagnetization  coefficient 
line  OB  are  proportionately  reduced.  This  results  in  an  attenuation  of  the  recorded  signal. 
It  may  be  observed  in  Figs.  2  and  3  that  the  degree  of  magnetization  of  an  element  which 
has  been  biased  but  is  otherwise  unrecorded  is  substantially  the  same  in  either  process,  and 
also  that  the  undistorted  operating  range  is  nearly  identical.  Because  of  these  conditions 
the  background  noise  and  dynamic  range  in  either  process  are  substantially  the  same,  and 
thus  when  operating  with  a  d-c  bias  there  is  little  advantage  in  providing  extra  equipment 
to  demagnetize  the  medium. 

It  is  now  customary  to  erase  by  demagnetization  and  employ  a  high-frequency  (super- 
sonic) biasing  current  during  recording.  The  action  of  the  high-frequency  bias  has  been 
variously  explained.  It  is  sufficient  to  say  here  that  it  serves  the  purpose  of  straightening 
out  the  bend  in  the  normal  magnetization  curve  around  the  origin.  The  resulting  action 
is  as  shown  in  Fig.  4.  The  recording  signal  may  then  operate  about  the  origin  0  and  along 
the  entire  straight  portion  of  the  curve  between  D  and  F.  The  remainder  of  the  process 
may  be  considered  to  be  similar  to  that  described  for  recording  with  a  d-c  bias  on  a  demag- 
netized medium  except  that  the  recording  range  is  doubled.  This  represents  a  doubling  of 
the  reproduced  signal  without  an  equivalent  increase  in  background  noise. 

There  is  an  even  greater  increase  in  signal-to-noise  ratio  inasmuch  as  the  elimination  of 
the  d-c  biasing  component  of  magnetization  serves  to  reduce  the  observable  background 
noise.  The  biasing  frequency  employed  in  this  method  of  recording  is  customarily  quite 
high  and  usually  leaves  no  permanent  impression  on  the  magnetic  medium.  Wherever  a 
biasing  frequency  is  recorded,  it  is  beyond  the  range  of  the  playback  equipment  and  is  not 
reproduced. 

In  common  with  all  methods  of  sound  recording,  magnetic  recording  introduces  a  certain 
amount  of  distortion  although  modern  magnetic  recording  systems  are  considerably  im- 
proved in  this  respect.  Distortion  is  introduced  during  the  recording  process  from  several 
sources.  The  most  common  source  of  distortion  arises  from  the  non-linear  properties  of 
the  recording  medium.  It  is  apparent  from  Figs.  2  and  3  that  there  is  an  optimum  value 
of  d-c  bias  which  will  minimize  curvature  distortion.  When  using  an  a-c  recording  bias 
there  is  a  minimum  value  which  is  sufficient  to  eliminate  the  curvature  of  the  normal 
magnetization  curve  around  the  origin.  Higher  values  than  the  necessary  minimum  may 
be  used,  but  there  is  no  advantage  to  be  obtained.  Distortion  resulting  from  intermodula- 
tion  between  the  bias  frequency  and  the  signal  frequencies  and  their  harmonics  may  be 


ERASING,  RECORDING^  REPRODUCING  PROCESSES      13-33 

kept  outside  the  operating  range  by  using  a  sufficiently  high  biasing  frequency.  When 
recording  with  an  a-c  bias,  distortion  is  increased  by  the  presence  of  a  residual  d-c  magnet- 
ization which  may  be  present  either  in  the  recording  and  reproducing  heads  or  in  the 
erased  medium. 

A  distortion  may  also  be  introduced  by  the  recording  head  if  the  magnetizing  field  at 
the  head  is  not  sufficiently  concentrated.  At  higher  frequencies  it  is  then  possible  for 
the  recording  signal  to  change  while  an  element  is  within  the  influence  of  the  head,  and 
the  Clement  will  retain  an  impression  of  the  strongest  magnetizing  force  to  which  it  is 
subjected.  The  use  of  narrow  recording  gaps  and  modern  high-permeability  pole  pieces 
has  greatly  reduced  this  type  of  distortion. 

REPRODUCTION.  In  the  reproducing  process  the  varying  magnetization  impressed 
on  the  medium  during  recording  produces  a  corresponding  flux  in  the  reproducing  head.  A 
portion  of  this  flux  threads  the  coil,  and  its  variation  due  to  the  motion  of  the  medium 
induces  the  signal  voltage.  Signal  voltages  are  normally  quite  low,  and  considerable 
amplification  is  required. 

The  process  by  which  the  reproducing  head  picks  up  the  signal  flux  is  essentially  similar 
for  any  of  the  heads  of  Fig.  1.  The  return  path  for  all  the  signal  flux  in  a  magnetized 
element  approaching  the  reproducing  head  is  through  the  air.  The  magnetic  state  of  such 
an  element  is  described  in  Figs.  2,  3,  and  4  by  a  point  on  an  open-circuit  demagnetization 
coefficient  line  such  as  OA.  As  the  element  approaches  the  head  some  of  the  flux  which 
leaves  the  medium  passes  to  the  head.  In  the  case  of  the  head  of  Fig.  l(a)  a  portion  of 
this  flux  threads  the  coil.  The  percentage  of  flux  which  threads  the  coil  increases  very 
rapidly  as  the  element  comes  directly  under  the  thin  lamination  and  decreases  as  rapidly 
thereafter.  Thus  the  total  flux  threading  the  coil  is  contributed  by  the  magnetized  ele- 
ments under  the  thin  lamination  and  those  immediately  adjacent.  In  the  case  of  the 
offset  pole  pieces  of  Fig.  1(6)  the  major  contribution  to  the  flux  threading  the  coil  is  made 
by  the  elements  in  the  immediate  region  of  the  overlap  point.  Elements  in  contact  with 
one  pole  tip  and  remote  from  the  overlap  point  are  either  insufficiently  coupled  or,  at  short 
recorded  wavelengths,  are  short-circuited  by  the  pole  tip  and  therefore  do  not  contribute 
to  the  flux  threading  the  coil.  The  ring-shape  head  of  Fig.  l(c)  operates  in  a  somewhat 
similar  manner.  The  head  serves  as  a  return  path  for  flux  leaving  the  surface  of  the 
longitudinally  recorded  medium.  With  the  exception  of  flux  from  elements  of  the  medium 
in  the  immediate  vicinity  of  the  air  gap,  the  return  path  does  not  include  the  reproducing 
coil.  Thus  the  flux  which  does  thread  the  coil  may  be  considered  as  being  contributed  by 
the  elements  in  the  immediate  region  of  the  air  gap. 

When  the  recorded  element  is  in  position  at  the  reproducing  point  its  magnetic  state  is 
described  in  Figs.  2,  3,  and  4  by  a  point  on  the  closed-circuit  demagnetization  coefficient 
line  OB.  The  slope  of  this  line  is  determined  by  the  total  reluctance  of  the  return  path 
for  the  element  in  question,  and  this  reluctance  is  affected  by  the  amount  of  contact 
between  the  medium  and  pole  pieces.  The  flux  threading  the  coil  is  then  equal  to  the  prod- 
uct of  the  Bi  intercept  of  the  point  on  OB,  the  normal  cross-sectional  area  of  the  contributr- 
ing  elements,  and  an  appropriate  leakage  factor.  The  contact  reluctance  of  the  narrow 
pole-tip  structure  of  Fig.  l(a)  is  higher  than  that  of  the  broad  pole-tip  structures  of  Figs. 
1(6)  and  l(c).  Therefore  the  slope  of  the  closed-circuit  demagnetization  line  OB  will  be 
less  in  the  former  case  and  the  flux  threading  the  coil  for  a  given  magnetization  of  the 
medium  will  also  be  less. 

The  open-circuit  signal  voltage  at  the  reproducing  head  is  proportional  to  the  rate  of 
change  of  flux  threading  the  coil.  This  factor  in  itself  is  responsible  for  a  6  db  per  octave 
rise  in  the  signal  characteristic.  There  are,  however,  several  other  factors  that  enter 
into  the  overall  frequency  response.  (1)  In  both  recording  and  reproduction  the  eddy- 
current  characteristics  of  the  head  may  introduce  some  attenuation  at  higher  frequencies. 
(2)  Following  recording,  the  medium  demagnetizes  to  a  degree  which  is  a  function  of  the 
length  of  the  recorded  magnets.  In  longitudinal  recording  the  resultant  attenuation  is  an 
inverse  function  of  the  recorded  wavelength.  (3)  In  reproduction  the  response  falls  off 
when  the  dimension  of  the  recorded  wavelength  becomes  comparable  to  the  active  region, 
of  the  reproducing  head.  Although  the  region  in  which  a  magnetized  element  of  the 
medium  may  cause  flux  to  thread  the  coil  is  not  sharply  defined,  an  effective  dimension 
may  usually  be  assigned,  and  this  dimension  has  been  called  the  "slit  width"  because  of 
its  similarity  to  the  optical  effect.  In  the  ring-shape  recording  head  experience  has  shown 
that  the  slit  width  is  from  10  to  40  per  cent  greater  than  the  air-gap  length.  (4)  In  certain 
reproducing  heads,  such  as  that  of  Fig.  1(6),  the  construction  is  such  that  the  pole  pieces 
are  somewhat  active  over  their  entire  dimension.  Thus  there  is  in  effect  a  secondary  slit 
width  equivalent  to  the  overall  dimension  of  the  poles  which  is  responsible  for  irregularities 
in  the  response  at  longer  wavelengths. 


13-34  ELECTROMECHANICAL-ACOUSTIC  DEVICES 


Lubeck  has  given,  for  ring-shape  heads,  a  general  expression  for  the  open-circuit  output 
voltage 

CD 


— 
X 


where  E  and  I  are,  respectively,  the  open-circuit  output  voltage  and  input  current, 
C  =  a  constant  for  the  particular  system,  N  -  number  of  turns  in  reproducing  coil, 
T>  -  linear  speed  of  medium,  s  =  effective  "slit  width"  of  reproducing  header  =  a  demag- 
netization constant  for  the  medium,  X  =  v/f  —  wavelength  of  recorded  signal,  /  «  fre- 
quency, and  w  =  27r/.  The  (electric  to  magnetic)  frequency  characteristics  of  the  record- 
ing and  reproducing  head  have  been  neglected  in  this  expression. 

Figure  5,  curve  (1),  shows  an  illustrative  frequency  response  characteristic.  The  factors 
determining  the  characteristic  are  independently  plotted.  It  is  seen  that  the  function 
sin  TS/X  which  is  the  "slit  width"  characteristic  plotted  in  curve  (3)  rises  to  a  maximum 


I 


(3)  THE  CHARACTERISTIC  6-DB  PER 
-    OCTAVE  RISE  AND  EFFECT  OF 
"SLIT  WtDTH*OF  PLAYBACK  HEAD- 

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(2)  EFFE 
DEMA6NE 
OF  M 

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mZATl 
EDIUM 

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(0  RESPONSE  AT 
PLAYBACK  HEAD" 

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VELOCITY 
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=  24  INCHES 
SECOND 

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S  =O.OO2  INCH 
T=O.O04-  INCH 

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0,4       Q.6    0.8    J 
FREQUENCY 


2  46 

IN  KILOCYCLES  PER  SECOND 


8      10 


20         30 


FIG.  5.    Frequency  Response  of  a  Longitudinal  Recording.    Open-circuit  playback  voltage  for  constant 

recording  current. 

when  the  recorded  wavelength  is  twice  the  effective  slit  width.  Thereafter  it  goes  through 
a  series  of  dips  and  peaks.  In  a  practical  head  design  where  the  working  gap  is  very  small, 
the  second  and  succeeding  peaks  are  not  reproduced  because  they  lie  in  the  region  where 
the  demagnetization,  curve  (2),  of  the  medium  causes  a  high  loss  and  they  usually  fall 
outside  the  frequency  range  of  associated  electrical  circuits. 

At  recording  speeds  in  current  use  the  slit  width  may  be  made  sufficiently  gtyia-U  so  that 
sin  3rs/X  can  be  replaced  by  the  angle  itself  within  the  working  frequency  range.  Under 
these  conditions 


E 


(2) 


and  it  is  seen  that,  except  for  demagnetization,  the  output  voltage  rises  6  db  per  octave 
and  is  not  influenced  by  the  recording  or  reproducing  speed  or  the  width  of  the  effective 
slit. 

It  is  usually  necessary  to  introduce  an  electrical  equalization  in  the  associated  recording- 
reproducing  circuits  in  order  to  obtain  the  desired  overall  frequency  response.  A  major 
portion  of  the  equalization  is  assigned  to  the  reproducing  circuits  where  it  is  limited  only 
by  the  frequency  spectrum  of  background  noise. 

When  due  consideration  is  given  to  the  overload  characteristic  of  the  recording  medium 
and  the  energy  distribution  of  the  recording  signal,  some  of  the  equalization,  may  be 
introduced  in  the  recording  circuits. 

SOURCES  OF  ICOISE.  Background  noise  arises  from  a  number  of  sources,  many  of 
wfaich  may  be  minimized  in  careful  design.  Among  such  sources  may  be  listed  amplifier 
inim*  thermal  noise,  external  noise  pickup  in  the  reproducing  head,  crosstalk  transferred 
from  &  strong  signal  on  adjacent  turns  of  the  medium  when  wound  on  the  storage  spool, 
BQfteohanieal  irregularities  on  the  surface  of  the  recording  medium  which  affect  the  degree 


RECORDING  MEDIA  13-35 

of  contact  with  the  reproducing  head,  variations  in  the  magnetic  properties  of  the  medium, 
and  variations  in  the  cross-sectional  area  of  the  medium.  The  remaining,  and  basic,  noise 
source  appears  to  be  caused  by  a  random  distribution  of  small  magnetic  inhomogeneities. 
Some  measurements  at  the  reproducing  head  of  noise  from  this  source  have  indicated  a 
fairly  uniform  frequency  distribution  of  noise  voltage  per  cycle.  In  most  cases,  after 
equalization,  this  background  noise  has  a  smooth  character  and  so  is  not  particularly 
disturbing.  Basic  noise  is  a  function  of  the  degree  of  magnetisation  of  the  medium.  It 
is  very  low  for  a  medium  which  has  been  thoroughly  erased  by  a-c  demagnetization,  but 
it  may  be  increased  to  a  considerable  extent  by  the  presence  of  the  a-c  biasing  flux.  The 
noise  level  may  be  considerably  increased  by  the  use  of  a  reproducing  head  in  which  there 
is  some  remanent  magnetization,  and  the  use  of  a  d-c  bias  increases  the  noise  markedly. 
Wherever  the  noise  is  decidedly  affected  by  the  presence  of  recording  bias,  there  is  detect- 
able, back  of  the  signal,  a  noise  which  rises  and  falls  with  the  magnitude  of  the  signal.  A 
certain  amount  of  such  noise  is  not  objectionable  as  it  is  partially  masked  by  the  signal. 

22.  RECORDING  MEDIA 

PHYSICAL  FORMS.  Most  of  the  early  magnetic  recording  equipment  employed  a 
solid  homogeneous  steel  wire  or  tape  as  the  recording  medium.  Recent  refinements  in  the 
art  of  electroplating  magnetic  alloys  and  developments  in  the  use  of  magnetic-powder- 
coated  paper  and  plastic  materials  have  removed  many  of  the  restrictions  on  the  form  of 
the  recording  medium.  Although  there  are  some  technical  differences  between  the  media, 
the  choice  of  physical  form  in  general  is  determined  by  the  desired  application. 

Magnetic  wire  is  generally  used  wherever  a  very  long  playing  time  is  required  or  a  com- 
pact equipment  is  necessary.  The  wire  diameter  is  limited  only  by  the  required  breaking 
strength,  and  diameters  between  0.004  in.  and  0.006  in.  are  most  common.  Splicing  is 
accomplished  by  tying  a  square  knot  which  passes  over  a  properly  designed  head  without 
difficulty.  A  longitudinal  recording  head  is  usually  employed,  and  the  wire  rides  in  a 
groove  which  is  worn  in  the  head.  This  method  of  recording  is  designed  to  distribute  the 
recording  signal  about  the  circumference  of  the  wire  and  serve  as  a  protection  against 
variation  of  the  reproduced  signal  as  the  wire  twists.  There  is,  however,  a  considerable 
variation  in  the  strength  of  the  recorded  signal  around  the  circumference  of  the  wire, 
particularly  at  short  wavelengths,  and  it  is  fortunate  that  smooth,  round  wire  does  not 
evidence  a  random  twisting  when  properly  used.  The  chief  disadvantage  of  the  use  of 
wire  is  the  tendency  to  uncoil,  snarl,  and  kink  whenever  a  free  end  becomes  loose.  For 
this  reason  it  is  advisable  to  mount  the  wire  in  a  magazine  containing  the  supply  and  take 
up  reels  and  remove  the  magazine  as  a  unit  wherever  record  storage  is  required. 

Magnetic  tape,  either  solid  or  plated,  is  most  useful  for  applications  requiring  a  con- 
tinuous record-reproduce-erase  cycle.  In  such  applications  the  tape  is  prepared  in  the 
form  of  an  endless  belt  which  is  spliced,  at  the  junction  point,  with  a  butt  weld.  When- 
ever a  more  extended  playing  time  is  required  the  tape  is  reeled  like  motion-picture  film. 
Recording  equipment  has  been  commercially  produced,  using  0.002  in.  by  0.050  in.  tape 
and  the  offset  recording  and  reproducing  pole  pieces  of  Fig.  1(5)  in  which  an  extended 
frequency  response  and  dynamic  range  are  realized  at  a  comparatively  low  operating 
speed.  The  solid  magnetic  recording  materials,  both  tape  and  wire,  have  the  common  dis- 
advantage that  accurate  dimensional  tolerances,  smooth  surface  conditions,  and  uni- 
formity of  heat  treatment  must  be  maintained  on  large  quantities  of  a  material  that  is 
inherently  very  difficult  to  fabricate. 

Coated  paper  and  plastic  recording  media  are  applicable  to  all  uses  where  requirements 
on  playing  time  are  not  excessive  and  its  low  remanence  can  be  tolerated.  It  has  been 
used  in  the  form  of  tape,  sheets,  and  disks.  The  material  is  cheap  to  manufacture  and 
may  be  cut  and  spliced  by  cementing.  The  material  is  usually  0.0015  to  0.0025  in.  thick. 
In  contrast  to  steel  wire  and  tape,  when  a  reel  of  the  material  is  removed  from  a  machine, 
there  is  little  danger  of  freely  uncoiling,  a  characteristic  that  is  very  desirable  in  the  home 
recording  field.  When  used  with  a  properly  designed  ring-type  recording-reproduciiig 
head  a  favorable  frequency  characteristic  may  be  obtained  with  a  comparatively  low 
operating  speed.  Output  levels  are  much  lower  than  can  be  attained  with  solid  wires  and 
tapes,  and  care  is  required  in  shielding  the  reproducing  head  and  in  the  design  of  reproduc- 
ing amplifiers.  Although  the  inherent  dynamic  range  of  the  coated  media  is  quite  large, 
thermal  and  hum  noise  levels  in  reproducing  equipment  may  limit  the  practical  dynamic 
range. 

Certain,  special  applications  of  magnetic  recording  have  employed  magnetic  media  in 
the  form  of  solid  disks  or  cylinders.  Most  frequently  the  recording  surface  is  electroplated 
on  a  non-ferrous  backing  material. 


13-36  ELECTROMECHANICAL-ACOUSTIC  DEVICES 

MAGNETIC  MATERIALS.  A  carbon-steel  recording  medium  was  used  in  most  of 
the  early  equipment  because  it  is  commercially  available  in  the  form  of  music  wire.  It 
has  very  little  value  in.  present-day  recording  applications.  The  coercive  force  is  low, 
approximately  40  oersteds,  and  thus  self-demagnetization  is  very  pronounced.  The 
residual  induction  is  approximately  10,000  gausses.  Because  of  demagnetization  the  short- 
wavelength  (high-frequency)  response  from  a  longitudinal  recording  is  very  limited.  The 
material  magnetizes  easily  from  the  recording  on  adjacent  turns  of  the  storage  spool,  and 
crosstalk  is  therefore  high.  The  basic  background  noise  is  rather  high  and  is  increased 
by  surface  corrosion  of  the  material. 

Thirteen  per  cent  chrome  steel  has  been  used  to  a  considerable  extent  in  both  magnetic 
wire  and  tape  recorders  because  of  its  resistance  to  corrosion.  This  is  a  ferritic  material 
depending  mostly  on  the  presence  of  carbon  for  its  magnetic  properties.  The  chromium 
is,  however,  instrumental  in  obtaining  the  desired  hardness.  When  properly  quenched, 
the  material  has  a  coercive  force  of  50-60  oersteds  and  a  residual  induction  of  7000- 
10,000  gausses.  A  frequency  response  and  dynamic  range  superior  to  carbon  steel 
may  be  obtained  from  the  material.  High  operating  speeds  are  required,  however,  when 
an  extended  frequency  range  is  desired.  To  prevent  formation  of  chromium  oxides,  which 
are  very  abrasive,  the  material  is  heat  treated  in  an  atmosphere  free  of  oxygen. 

Nickel-chromium  stainless  steels  such  as  18  per  cent  chromium,  8  per  cent  nickel  have 
been  used  in  magnetic  wire  recorders.  This  is  an  austenitic  steel  which  may  be  hardened 
by  cold  working.  A  somewhat  more  uniform  product  is  obtained  when  the  cold-worked 
material  is  then  age  hardened.  After  such  treatment  the  coercive  force  is  of  the  order 
of  150  to  350  oersteds  with  a  possible  maximum  residual  induction  of  7000  gausses. 

Several  other  magnetic  alloys  possess  properties  which  make  them  exceptionally  satis- 
factory as  magnetic  recording  media.  One  such  material  is  known  as  Vicalloy,  a  workable 
permanent-magnet  alloy  which  is  heat  treated  to  obtain  the  desired  magnetic  properties. 
A  typical  Vicalloy  composition  is  38  per  cent  iron,  52  per  cent  cobalt,  and  10  per  cent 
vanadium.  Vicalloy  recording  tape  has  been  commercially  produced  with  a  coercive  force 
of  225  oersteds  and  a  residual  induction  of  6000  gausses. 

One  of  the  chief  disadvantages  of  the  solid  magnetic  recording  media  has  been  the 
difficulty  of  maintaining  a  uniform  recording  sensitivity  and  background-noise  level 
throughout  the  entire  length  of  the  medium.  Recently,  an  electroplated  medium  has  been 
developed  in  order  to  correct  this  disadvantage  and  at  the  same  time  provide  a  cheaper 
recording  material.  The  recording  surface  consists  of  a  thin  layer  (approximately  0.0003 
in.)  of  a  nickel-cobalt  alloy  plated  on  hard  brass  wire  or  tape.  The  magnetic  and  physical 
properties  are  controlled  in  the  plating  process,  and  subsequent  working  or  heat  treatment 
is  not  required.  The  coercive  force  is  approximately  200  oersteds  with  a  residual  induction 
of  the  order  of  8000  gausses.  A  satisfactory  frequency  range  may  be  obtained  from  this 
material  at  a  comparatively  low  operating  speed. 

The  use  of  powdered  magnetic  materials  applied  to  paper  or  plastic  carriers  has  recently 
received  considerable  attention  in  this  country.  Such  a  recording  medium  is  comparatively 
cheap  to  manufacture  and  may  be  produced  with  very  uniform  and  stable  properties. 
Various  magnetic  materials  in  powder  form  are  being  investigated  for  properties  advan- 
tageous to  magnetic  recording.  One  such  material,  black  magnetic  iron  oxide,  is  com- 
mercially available  in  the  required  finely  divided  form.  The  powdered  material  is  nor- 
mally dispersed  in  a  plasticized  lacquer  and  applied  to  the  carrier  to  a  thickness  of  approxi- 
mately 0.0005  in.  When  the  powdered  magnetic  material  is  very  finely  divided  and  uni- 
formly dispersed  in  the  binder  very  excellent  results  have  been  obtained  both  in  frequency 
response  and  background  noise.  Coercive  forces  ranging  between  100  and  500  oersteds 
have  been  measured.  The  residual  induction  is  very  low,  and  of  the  order  of  a  few  hundred 
gausses.  Demagnetization  of  signals  of  short  wavelengths  appears  to  be  less  pronounced 
in  the  powdered  materials  than  in  solid  materials  of  equal  coercive  force.  Output  levels 
are  very  low,  and,  unless  a  very  wide  sound  track  is  used,  unusual  precautions  are  required 
in  the  design  of  reproducing  equipment  which  will  take  full  advantage  of  the  inherent 
dynamic  range  of  the  medium. 

In  Germany  considerable  work  has  been  done  on  tape  in  which  the  magnetic  medium 
is  ferric  and  ferrous  oxide  in  individual  particles  about  1  micron  in  size.  This  material  is 
manufactured  from  precipitated  finely  divided  black  magnetic  iron  oxide  by  further 
oxidization  in  an  agitated  drier.  The  red  ferric  oxide  has  the  crystal  structure  of  the  mag- 
netic oxide  and  is  also  magnetic.  Several  types  of  recording  tape  have  been  manufactured 
in  which  the  magnetic  oxides  are  either  cast  on  the  surface  of  a  plastic  carrier  such  as 
cellulose  acetate  or  polyvinyl  chloride  or  are  dispersed  throughout  a  tape  of  polyvinyl 
chloride  in  a  50-50  mixture.  It  is  reported  that  a  frequency  response  uniform  to  within 
d=2  db  from  50  to  10tOOO  cps  is  obtained  at  a  tape  speed  of  30  in.  per  sec,  the  overall  noise 
level  is  very  low,  and  the  useful  life  of  the  recorded  tape  exceptionally  high. 


RECORDING  INSTRUMENTS  13-37 


BIBLIOGRAPHY 

V.  Poulsen,  Ann.  d.  Pkysik,  VoL  3,  754  (Dec.  13,  1900). 

V-  Poulsen,  Electrician,  Vol.  46,  208  (Nov.  30,  1900) 

L.  Bdlstab,  E.T.Z.,  VoL  22,  57  (Jan.  17,  1901) 

J.  H.  West,  E.T.Z.,  VoL  22,  181  (Feb.  21,  1901). 

A.  Nasarischwily,  E.T.Z.,  Vol.  42,  1068  (Sept.  22,  1921). 

W\u*    30ril5|7and  Glelm  W"  Carpenter'  U-  S-  Pat"  1.640.881,  appHed  for  March  26,  1921;  issued 

C/Stilie,  E.T.Z',  VoL  51,  449  (March  27,  1930). 

E.  Meyer,  Handbuch  der  experimental  Pkysik,  VoL  XVII,  Chapter  11. 

C.  N.  Hickman,  Bell  Lab.  Rec.,  VoL  11,  308  (June  1933). 

R.  F.  Mallina,  Bell  Lab.  Rec.,  VoL  13,  200  (March  1935). 

E.  Schuller,  E.T.Z. ,  VoL  56,  1219  (Nov.  7,  1935). 

H.  Lubeck,  Akustiche  Zeit.,  VoL  2,  273  (November  1937). 

C.  N.  Hickman,  Bell  Sys.  Tech.  J.,  VoL  16,  165  (April  1937). 
S.  J.  Begun,  J.  S.M.P.E.,  VoL  29,  216  (August  1937). 

A.  E.  Barrett  and  C.  J.  F.  Tweed,  J.  I.E.E.,  VoL  82,  265  (March  1938). 

D.  E.  Wooldridge,  U.  S.  Pat.  2,235,132,  applied  for  July  29,  1939;  issued  March  18,  1941. 
M.  Camras,  U.  S.  Pat.  2,351,004,  applied  for  Dec.  22,  1941;  issued  June  13,  1944. 

S.  J.  Begun,  Proc.  I.R.E.,  VoL  29,  423  (August  1941). 

Marvin  Camras,  Radio  News,  Radionics  Dept.,  VoL  1,  3  (November  1943). 

Hershel  Toomim  and  David  Wildfeuer,  Proc.  I.R.E.,  Vol.  32,  664  (November  1944). 

L.  C.  Holmes  and  D.  L.  Clark,  Electronics,  VoL  18,  126  (July  1945). 

S.  J.  Begun,  Communications,  VoL  26,  31  (April  1946). 

D.  E.  Wooldridge,  Trans.  A.I.E.E.,  VoL  65,  343  (June  1946). 

C.  W.  Hansell  and  others,  Dept.  of  Commerce,  Office  of  the  Publication  Board,  PB  1346. 

J.  Z.  Menard,  Dept.  of  Commerce,  Office  of  the  Publication  Board,  PB  12659. 


MECHANICAL  RECORDING 
AND  REPRODUCING  OF  SOUND 

By  L.  Vieth  and  H.  A.  Henning 

The  requirements  for  high-fidelity  recording  and  reproduction  of  sound  are  (a)  that  the 
whole  system,  from  the  point  where  the  sound  reaches  the  pickup  device  to  the  point  where 
it  is  actually  reproduced  as  sound,  shall  have  a  linear  relationship  between  its  input  and 
its  output,  (6)  that  the  system  have  a  uniform  response  versus  frequency  characteristic, 
since  any  complex  wave  may  be  resolved  into  the  sum  of  simple  sinusoidal  terms,  and  (c) 
that  the  system  have  a  linear  relationship  between  its  phase  shift  and  the  frequency  im- 
pressed upon  it  and  that  the  phase  angle  have  a  value  of  ±mr  at  zero  frequency  where 
n  —  0,  1,  2,  3,  etc.  The  requirements  of  individual  components  may  vary  from  the  require- 
ments of  uniform  response  versus  frequency  characteristic  (&),  but  in  general  the  re- 
quirements for  the  system  apply  to  its  components. 

Most  sound  recording  and  reproducing  systems  and  components  represent  a  compromise 
between  the  requirements  for  high-quality  reproduction  and  the  commercial  requirements 
of  size,  cost,  and  general  adaptability  for  a  specific  use.  These  compromises  have  resulted 
in  a  wide  range  of  overall  performances.  Space  limitations  prohibit  discussion  of  more 
than  a  few  of  the  typical  instruments  in  commercial  use  in  the  mechanical  recording  and 
reproducing  of  sound. 

23.  RECORDING  INSTRUMENTS 

Mechanical  recording  is  used  almost  exclusively  in  the  present-day  phonograph  industry, 
in  electrical  transcriptions  for  broadcast  purposes,  and  in  stenographic  applications 
(dictation  machines)  and  general  utility  recorders.  Most  mechanical  recording  is  done  on 
disks  varying  in  diameter  from  6  in.  for  general  utility  equipment  to  16  in.  in  broadcast 
transcriptions.  There  are,  in  addition,  several  commercial  devices  in  which  the  recording 
medium  takes  the  form  of  standard  motion-picture  film  base.  The  materials  of  which 
these  media  are  made  are  discussed  elsewhere.  They  may  be  grouped,  in  general,  into  two 
classes:  (a)  materials  for  recording  for  instantaneous  playback;  (Z>)  materials  for  recording 
for  subsequent  processing.  However,  the  function  of  the  recording  instrument  is  the  same 
in  all :  to  transfer  to  the  recording  medium,  in  the  form  of  emboesed  or  engraved  modula- 
tions, a  counterpart  of  the  voltage  impressed  on  the  recorder  modified  in  frequency  char- 
acteristic in  conformance  with  some  preconceived  plan  of  equalization  which  affords  the 
optimum  use  of  the  recording  equipment. 

A  typical  recording  instrument  used  in  this  work  is  illustrated  by  Fig,  1,  which  shows  a 
phantom  view  of  a  lateral-type  recorder  (one  in  which  the  stylus  moves  laterally  parallel 


13-38  EUBCTROMECHAlSriCAL-ACOITSTIC   DEVICES 


Magnetizing 
Winding 


Magnet 


stylus 


f$FiG.  1.     Electromechanical  Type  Recorder 


s:      _ 


-* 


S  £«>  c/ig  ^g  >•  cng  Q  •££ -S 

O  u11-     Bog     «C  CO  ^C  (a  -1"   -1 

3  |«J^~-0=  »  .£=  ^  ^gfe 

P  c;        oo^5o  "5  oo  o  333 

.<  ~Z^o  OO  CQO  O  OO  O  Q£c/3  OS 


FIG.  2.     Equivalent  Circuits  of  EJlectromechanical  Recorder  Shown  in  Fig.  1 


vo  Velocity  In  dl 
5  o  S  < 

"^n 

N 

*-— 

*^» 

^- 

K—  2O 

0    50      10O               500    1QOO             5OOO    10,OOO 

Frequency  in  Qyctes  per  Second 
Response  TS.  Fretniency  Characteristio  of  a  Lateral  Type  Recorder 


RECORDING  INSTRUMENTS 


13-39 


to  the  radius  of  the  disk)  designed  by  Bell  Telephone  Laboratories*some  years  ago.  The 
same  general  structure  has  been  used  in  vertical-type  recorders  (one  in  which  the  stylus 
moves  normal  to  the  face  of  the  disk) .  The  structure  is  a  mechanical  filter  whose  electrical 
equivalent  is  shown  in  Fig,  2,  in  which  the  current  in  the  second  mesh  is  analogous  to  the 


Fia.  4.    Cross-sectional  View  of  Vibrating  System  and  Associated  Magnetic  Circuit  of  Western  Electric 

Co.  1A  Feedback  Recorder 


stylus  velocity.  Figure  3  shows  a  typical  response  versus  frequency  characteristic  of  such  a 
recorder.  The  loss  at  the  lower  frequencies  is  part  of  a  preconceived  equalization  plan 
which  makes  the  best  use  of  the  recording  medium  by  limiting  amplitudes  in  the  interest 
of  record  space  economy. 

The  vibratory  system  of  a  more  modern  version  of  recording  instrument  is  shown  in 
Fig.  4.  This  device,  also  designed  at  Bell  Telephone  Laboratories,  utilizes  the  principle 
of  stabilized  feedback  to  control  the  stylus  velocity  and  involves  an  associated  amplifier 
in  which?  the  recorder  becomes  an  actual  transmission  element  as  well  as  a  terminal  trans- 
ducer. Schematically,  the  device  associated 
with  an  appropriate  amplifier  may  be  repre- 
sented as  shown  in  Fig.  5.  The  output  volt- 
age E<i  of  the  amplifier  is  supplied  to  the  driv- 
ing coil  of  the  recorder,  thereby  driving  the 
stylus  with  a  velocity  V.  Motion  of  the 
stylus  in  turn  generates  in  a  suitable  generat- 
ing element,  such  as  a  small  coil  moving  in  a 
magnetic  field,  the  voltage  E$.  This  voltage 
is  returned  to  the  amplifier  input  through  a 
control  circuit  which  may  be  either  passive 
or  active.  The  voltage  available  after  modt- 
fication  in  the  control  circuit  is  designated  E^. 
The  voltages  and  velocities  here  referred  to  are  to  be  considered  as  having  both  magnitude 
and  phase  and  hence  can  be  represented  in  complex  number  notation.  Then 


»,  ___  J 


FIG.  5.    Schematic  Representation  of  an.  Elec- 
tromechanical Feedback  System 


=  S 


(1) 


To  obtain  a  simple  expression  for  the  relation  of  the  stylus  velocity  V  to  the  signal 
voltage  E,  let 


and 


and  hence 


Et 
V 


V  E* 


(3) 
(4) 


13-40  ELECTROMECHANICAL-ACOUSTIC  DEVICES 


The  product  AB  thus  defines  the  transmission  around  the  loop  formed  by  the  amplifier, 
recorder,  and  feedback  control.  The  value  of  E4  from  this  equation  can  now  be  substituted 
in  the  relation  EI  —  E  +  E±  to  obtain 

(5) 


which,  together  with  eq.  (2),  gives 

V 


1  -  AB 


E  - 


(6) 


The  right-hand  side  of  eq.  (6)  is  the  familiar  expression  for  feedback  amplifiers  in  gen- 
eral, and  the  rules  for  stability,  gain,  distortion,  etc.,  are  equally  applicable.  In  particular, 
when  AB  is  very  large  compared  to  unity 


-B 


E        (\AB\  »  1) 


(7) 


which  indicates  that  over  the  frequency  range  considered  the  velocity  of  the  stylus  is 
independent  of  the  amplifier  gain  or  the  efficiency  of  the  recorder.  Variations  in  B, 
however,  directly  affect  the  performance,  and  hence,  if  a  fiat  frequency  response  is  desired, 
B  must  remain  constant.  However,  since  B  is  the  product  of  the  mechanical-electrical 
conversion  factor  E$/V  and  the  control  factor  EJEz,  it  will  be  seen  that  these  factors  may 
vary  as  long  as  their  product  remains  constant.  It  is  a  simple  matter  to  maintain  the 
factor  Ez/V  constant,  and  hence  a  fiat  response  characteristic  depends  only  upon  keeping 
the  control  factor  constant. 

If  eq.  (6)  is  rewritten  to  include  noise  and  distortion  products  as  well  as  signal,  it  becomes 


-  AB   '    1  -  AB   '    1  -  AB 


(8) 


where  n  and  d  are  the  noise  and  distortion,  respectively,  introduced  in  the  amplifier  and 
recorder  without  feedback.  Hence,  when  AB  is  large  compared  to  unity,  both  the  noise 
and  the  distortion  components  are  reduced  as  compared  with  the  corresponding  effects 

in  a  non-feedback  system. 
Variations  in  the  im- 
pedance of  the  recording 
medium  which  act  upon 
the  stylus  during  cutting 
may  be  regarded  as  noise 
or  distortion  introduced 
in  the  recorder,  and  their 
effect  upon  the  vibra- 
tional  velocity  is  also  re- 
duced by  the  above  factor. 
This  is  equivalent  to  a 
manifold  increase  in  the 
driving-point  impedance. 
A  frequency  response 
characteristic  with  and 
without  feedback  on  a 
typical  feedback-type  re- 
corder is  shown  in  Fig.  6. 
The  flat  frequency  re- 
sponse characteristic 

i-    ..   j         -     j.  -,  ,.        .         ,  shown   is,    of   course,    of 

hmited  use  m  disk  recording  since  the  large  amplitudes  at  low  frequencies  are  prohibitive. 
However,  since  the  overall  characteristic  of  the  device  is  flat,  pre-equalization  can  be 
accomplished  flexibly  and  simply  by  electrical  networks. 


RESPONSE  IN  DECIBELS 

JL  _  10  CJ  K  ,, 
O  O  O  O  0  o  o 

/*• 

A- 
WITHOUT 
FEEDBACK^ 

/ 

\ 
\ 

/ 

\ 

\ 

f* 

/ 

\ 

\ 

/ 

/ 

B-wrr 

H  FEE 

)BA 

V 
^ 

\ 

X, 

•7 

\ 

10,000  20/>00 


20         40    60       100       200      400  1000    2000    4000 

FREQUENCY  IN  CYCLES   PER  SECOND 

FIG.  6.     Curves  Showing  the  Stylus  Velocity  for  a  Constant  Signal  In- 
put to  the  Recorder  Amplifier  System.     (A)  Without  feedback;   (5) 
with  properly  controlled  feedback. 


24.  RECORDING  AND  REPRODUCING  MEDIA 

Two  broad  classifications  may  be  made  of  mechanical  recording  media:  (a)  the  "wax" 
disk  intended  primarily  for  subsequent  processing  and  duplication,  and  (6)  the  instantan- 
eous media  disk  or  otherwise,  intended  mainly  for  reproduction  directly  from  the  embossed 
or  engraved  master  The  latter  are  occasionally  used  for  subsequent  processing  and  dupli- 
<*'  cylinders  are  used  to  some  extent  in  dictating  machines  as  direct  playback 


EECOEDING  AND  REPRODUCING  MEDIA 


13-41 


FIG.  7.     Western  Electric  Co.  Lateral  Recording  Styliis 


RECORDING  DISKS  FOR  PROCESSING  AND  DUPLICATION.  The  shape  of  the 
recording  stylus  vanes  somewhat  in  commercial  practice  in  the  cutting  of  disks  for  subse- 
quent processing  and  duplication.  In  general  the  cross-section  and  particularly  the  curva- 
ture of  the  unmodulated  groove  are  kept  within  fairly  narrow  limits.  The  stylus  used  on 
Western  Electric  recorders  has  a  tip  radius  of  0.0021  in.  ±0.0001  in.  Its  contour  is  illus- 
trated in^  Fig.  7.  The  disposition  of  grooves  and  land  between  grooves  is  a  matter  of 
compromise  between  playing  time  and  signal-to-noise  ratio  and  other  factors.  For  lateral 
recording  the  groove  pitch  is  usually  between  0.007  and  0.010  in.  as  illustrated  by  Fig.  7. 
The  maximum  safe  amplitude  at  any  point  on  the  groove  will  depend  upon  the  amplitude 
and  phase  of  the  sound  recorded  on  the  adjacent  grooves.  For  a  groove  situated  between 
two  unmodulated  grooves,  the  maximum  amplitude  for  the  condition  pictured  in  Fig.  7 
will  be  nearly  0.004  in.  The  maximum  amplitude  for  two  adjacent  grooves  of  equal  ampli- 
tude and  180°  out  of  phase  with  each  other  will  be  slightly  less  than  0.002  in.  The  space 
between  grooves  may  be  compressed,  in  vertically  cut  recordings,  to  increase  the  playing 
time  of  a  record  with  less  sacrifice 
in  recording  level  than  for  lateral 
recording. 

In  cutting,  the  "wax'*  (more 
correctly  a  metallic  soap)  must  be 
leveled  on  the  recording  machine 
with  reasonable  care,  and  the 
stylus  must  be  sharp  and  so  ground 
that  the  cut  will  be  very  clean.  As 
it  is  cut  the  wax  shaving  is  removed 
by  air  suction.  The  operator  is 
aided  in  maintaining  the  correct 
depth  of  cut  by  a  so-called  advance 
ball.  Commonly  an  advance  ball 
is  a  ball-shaped  sapphire  mounted 
in  an  adjustable  holder  which  is  in 
turn  fastened  to  the  recorder.  The 
advance  ball  rides  lightly  over  the 
wax  close  to  the  stylus  and  serves  to  maintain  uniform  depth  of  cut  in  spite  of  small  in- 
accuracies of  leveling  of  the  "wax"  or  deviation  from  planeness.  The  advance  ball  is 
adjusted  relative  to  the  stylus  by  observing  the  cut  with  a  calibrated  microscope. 

In  commercial  cutting,  for  processing,  both  solid  and  flowed  waxes  are  popular.  Solid 
disks  are  shaved  with  a  sapphire  knife  to  a  highly  polished  surface  on  a  sturdy  high-speed 
turntable.  After  recording  and  processing  the  disk  is  reshaved  and  recut  until  it  becomes 
too  thin  to  be  used  with  safety.  The  flowed  disk  is  a  thin  layer  of  wax  flowed  onto  a  metal 
surface.  The  wax  layer  is  cut  once  and  discarded,  the  metal  backing  being  reused  indefi- 
nitely. Such  a  recording  medium  is  extremely  smooth,  homogeneous,  and  free  from 
mechanical  strains  incidental  to  shaving  the  "waxes"  by  the  older  method. 

DUPLICATION  OF  DISK  RECORDS.  The  surface  of  the  wax,  after  being  engraved, 
is  -rendered  electrically  conducting.  This  can  be  done  in  a  number  of  ways,  such  as  dusting 
with  graphite  and  bronze  or  other  electrically  conducting  powders,  by  the  chemical  pre- 
cipitation, of  silver,  .or  by  sputtering  a  suitable  material  such  as  gold  or  silver  in  a  cathode 
sputtering  chamber. 

The  wax  is  then  electroplated  with  copper,  sometimes  to  a  thickness  of  about  x/32  in-» 
or  sometimes  only  to  a  thickness  of  a  few  thousandths  of  an  inch  and  then  backed  up  by  a 
thicker  metal  plate  of  suitable  material.  This  electrodeposited  plate  is  then  a  negative  of 
the  wax  and  is  called  a  "master."  From  this  negative  or  master,  positive  copies  may  be 
made.  In  commercial  practice  the  copies  or  records  are  usually  pressed  of  some  thermo- 
plastic material.  If  the  sound  reproduction  from  a  few  records  carefully  pressed  from  a 
master  is  satisfactory,  duplicate  stampers  are  produced  in  order  that  the  master  may  be 
preserved.  These  duplicate  stampers  are  used  for  pressing  or  molding  the  commercial 
release  records. 

From  the  master  one  or  more  "mother"  plates  are  made  by  eJectrodepodition.  These 
mothers  are  positives  and  serve  as  the  forms  on  which  the  final  matrices  or  stampers  are 
deposited.  One  of  the  important  problems  in  this  process  is  the  treating  of  the  metal 
surfaces  of  the  masters  and  mothers  so  as  to  permit  subsequent  separation.  The  problem 
is  to  produce  a  surface  which  will  still  be  conducting,  but  which  is  not  "clean'*;  i.e.,  the 
deposited  metal  must  not  be  in  sufficiently  intimate  contact  with  the  metal  of  the  mold 
to  cause  permanent  adherence.  This  may  be  done  by  either  of  two  methods:  (a)  by  the 
application  of  a  thin  "mechanical"  layer  of  some  substance  such  as  grease,  graphite,  etc., 
or  (&)  by  a  chemical  film  usually  produced  by  treating  the  previously  cleaned  metal  surface 


13-42  ELECTROMECHANICAL-ACOUSTIC  DEVICES 

with  some  chemical  that  will  react  with  the  metal  and  produce  an  insoluble  compound. 
Typical  substances  for  the  purpose  are  soluble  sulfides  such  as  sodium  sulfide  (Na2S)  or 
the  yellow  "polysulfide"  (Na2Sa:).  These  will  react  with  the  metal  to  form  films  of  the 
corresponding  sulfides.  Frequently  the  phonograph  industry  has  employed  a  film  of 
silver  iodide,  formed  by  first  treating  the  copper  surface  with  a  silver  cyanide  solution. 
The  solution  is  either  poured  over  the  previously  cleaned  surface  or  mixed  with  an  inert 
substance  such  as  whiting  or  calcium  carbonate  (CaCOs)  to  form  a  paste  which  is  rubbed 
over  the  copper.  A  thin  film  is  thus  deposited  by  immersion.  The  film  is  then  treated 
with  a  weak  iodine  solution.  The  iodine  solution  acts  upon  the  silver  to  form  a  thin  film 
of  silver  iodide  upon  which  the  copper  can  then  be  deposited  and  subsequently  separated. 
Numerous  other  methods  are  employed  in  the  industry. 

Until  recently  the  records  themselves  were  molded  of  a  thermoplastic  mixture  of  shellac 
and  earth  fillers  and  were  generally  played  with  steel  needles.  The  fillers  were  purposely 
made  somewhat  abrasive  in  order  to  grind  the  needle  point  to  fit  the  groove  since  the 
needles  were  not  usually  accurately  shaped  to  a  suitable  contour.  The  pressures  at 
the  needle  point  of  a  typical  needle,  before  being  ground  to  shape,  are  extremely  high,  and 
record  wear  under  this  condition  is  serious.  In  more  recent  years,  reproducers  of  lower 
mechanical  impedance  have  become  available,  which  require  much  lower  pressures.  Accu- 
rately contoured  permanent  reproducing  points  provide  further  relief  by  reducing  vibra- 
tory mass  and  maintaining  a  definite  small  raolius  of  curvature.  Any  abrasive  in  the 
records  adds  to  the  background  noise,  hence  homogeneous,  non-abrasive  records  such  as 
those  made  of  cellulose  acetate  or  vinylite  are  coming  to  be  used.  Such  materials  are,  in 
fact,  used  almost  universally  in  transcription  recordings  for  broadcast  purposes. 

Records  of  the  older  shellac  mixtures  or  of  some  of  the  more  recently  developed  plastic 
materials  are  made  in  about  the  same  manner,  although  the  time,  temperature,  and  pres- 
sure cycles  may  differ  widely.  The  material  to  be  molded  is  usually  heated  to  a  suitable 
softening  temperature  (of  the  order  of  300  deg  fahr)  either  on  an  auxiliary  hot  plate  or  in 
the  record  press  itself  in  which  the  stampers  are  mounted.  The  stampers  are  commonly 
mounted  in  the  press  on  platens  which  are  heated  and  then  cooled  according  to  some  pre- 
determined cycle.  The  material  may  be  further  heated  in  the  press  at  a  suitable  tempera- 
ture before  the  pressure  of  the  press  is  applied  upon  it.  The  pressure  applied  is  of  the 
order  of  2000  Ib  per  sq  in.  for  many  plastic  materials.  This  pressure  is  maintained  while 
the  platens  are  cooled  for  a  suitable  time,  after  which  the  press  is  opened  and  the  record 
removed. 

INSTANTANEOUS  PLAYBACK  DISKS.  This  term  covers  a  multitude  of  plastics 
and  metals  which  lend  themselves  to  easy  engraving  or  embossing.  A  few  will  be  discussed. 
In  the  higher-quality  field  of  instantaneous  recording  media  the  12  in.  and  16  in.  lacquer- 
coated  disks  of  glass  or  metal  are  most  common.  A  variety  of  materials  may  be  com- 
pounded with  cellulose  nitrate  to  form  a  lacquer  that  is  easily  engraved.  These  coatings 
are  approximately  0.010  in.  thick  and  cut  cleanly  with  a  recording  stylus  similar  to  that 
used  for  cutting  "wax,"  although  it  is  general  practice  to  somewhat  dull  the  cutting  edge 
to  a  radius  of  the  order  of  0.0003  in.  The  shaving  may  be  directed  toward  the  center  and 
collected  about  the  center  pin  of  the  disk  or  it  may  be  removed  by  suction  apparatus. 

Disks  of  this  kind  are  sometimes  used  as  masters  for  subsequent  processing  and  duplica- 
tion in  the  same  manner  as  described  for  wax  disks.  When  the  disks  are  used  for  instan- 
taneous playback,  the  elastic  properties  of  the  recording  material  seriously  affect  the  re- 
sponse characteristic.  High-frequency  losses,  particularly  at  low  linear  velocities,  are 
very  severe  and  are  not  too  satisfactorily  relieved  by  predistortion  in  recording.  Depend- 
ing upon  the  type  of  reproducer  used  on  these  disks,  they  may  be  good  for  from  one  to 
fifty  playings. 

In  the  general  utility  field  recording  disks  are  most  commonly  made  of  vinylite  or 
vinylchloride.  Gelatin  and  soft  metals  such  as  aluminum  are  also  used.  These  materials, 
in  general,  are  not  suitable  for  engraving,  and  in  most  applications  an  embossing  stylus 
is  used  which  rubs  a  shallow  groove  into  the  recording  material,  modulating  the  groove  in 
accordance  with  the  signal  impressed  on  the  recorder.  The  contour  of  such  styli  varies 
widely.  Grooves  so  rubbed  are  about  0.001  in.  deep  and  the  maximum  amplitude  rarely 
exceeds  0.001  in.  Signal-to-noise  ratios  are  relatively  low.  These  applications  are  defi- 
nitely not  high  quality  in  their  present  state  of  development,  emphasis  being  placed  on 
obtaining  good  articulation.  Useful  playing  life  is  uncertain.  Extremely  light  reproducers 
and  soft  reproducing  styli  are  required  if  more  than  a  few  playings  are  anticipated.  Disks 
in  general  do  not  exceed  6  or  8  in.  in  diameter. 

RECORDING  MEDIA  OTHER  THAN  DISK.  Although  the  recording  medium  on 
most  general-utility  recording  machines  takes  the  form  of  a  disc,  two  exceptions  worthy 
of  Etote  are  made.  The  first  of  these  is  film  on  which  the  recorded  grooves  are  embossed 
in  iteiical  pattern  on  an  endless  loop  of  standard  35-mm  safety  motion-picture  film  base 


REPRODUCING  INSTRUMENTS 


13-43 


(Amertype  Recordgraph) .  The  other  exception  is  the  well-known  wax  cylinder  which 
approximates  in  its  consistency  the  wax  disk  used  in  recordings  for  processing.  Here  a 
helix  is  cut  in  the  cylinder  to  be  erased  for  subsequent  reuse  after  playing  (Dictaphone  and 
Hidiphone) . 

A  unique  combination  of  mechanical  recording  and  optical  reproduction  is  found  in  the 
Phinpps-Miller  system,  in  which  a  wide-angle  stylus  cuts  an  extremely  shallow  groove 
through  the  darkened  emulsion  of  a  photographic  film.  The  thin  emulsion  is  therefore 
cut  away  in  varying  widths  so  that  the  transparent  body  of  the  film  presents  a  pattern 
similar  to  that  of  variable  area  sound-on-film  recording.  Reproduction  is  accomplished 
optically  in  a  similar  manner  to  ordinary  sound  on  film. 


25.  REPRODUCING  INSTRUMENTS 

The  function  of  the  reproducer  in  a  mechanical  recording  system  is  to  transform  into 
electrical  energy  the  modulations  of  the  recorded  disk  or  duplicate  thereof.  Since  the 
inception  of  so-called  "electrical  recording"  in  contrast  to  the  older  method  of  acoustical 
recording,  magnetic-type  reproducers  have  played  an  important  part.  Such  reproducers 
have  taken  a  variety  of  forms,  but  in  general  an  armature  is  rigidly  attached  to  a  stylus 
holder.  The  armature,  moving  in  a  magnetic  field,  generates  a  voltage  in  associated  coils. 
Typical  examples  of  such  structures  are  illus- 
trated in  Figs.  8  and  9.  Both  these  devices  ^"" — "  ~""-\  Magnet 
and  others  of  the  same  vintage  are  lateral-type 
reproducers  and  are  intended  for  use  with  a  re- 
placeable steel  stylus  or  needle.  To  provide 
sufficient  rigidity  against  flexing,  such  styli  in 
themselves  are  quite  heavy,  and  their  mass  plus 
that  of  associated  driving  elements  produces  a 


FIG.  8.    Early  Type  Western  Electric  Co.  Oil-damped 
Lateral  Reproducer 


FIG.  9.     Early  Type  RCA  Victor  Re- 
producer 


high  driving-point  impedance  with  corresponding  record  wear  and  distortion.  In  general, 
the  same  applies  also  to  the  fitted  jewel  styli  used  in  conventional-type  lateral  reproducers, 

In  reproducing  vertical  recordings  the  driving  force  is  in  direct  line  with  the  stylus  axis 
and  flexural  rigidity  is  not  so  important  a  requirement.  The  mass  of  the  stylus  aad  asso- 
ciated vibratory  system  can  therefore  be  kept  comparatively  small,  thus  redseang  the 
driving-point  impedance  so  that  record  wear  is  greatly  reduced.  Vertical  reproducers  m 
general  have  been  built  around  the  moving-coil  principle  and  invariably  are  associ^ed 
with  a  jeweled  (diamond  or  sapphire)  stylus  that  has  been  polished  to  fit  t&e  recorded 
groove  accurately.  .  , 

One  of  the  more  common  of  reproducers  is  the  piezoelectric  crystal  type,  in  wiucn  a 
Rochelle  salt  crystal  element  is  secured  to  a  stylus  holder  at  one  end  and  anchored  at  tbe 
other.  Stylus  movement  causes  a  flexing  of  the  crystal  element,  which  produces  a  voltage. 
In  general  that  type  of  reproducer  is  of  fairly  high  driving-point  impedance,  although  some 
•of  the  higher-quality  devices  provide  a  distinct  improvement  in  this  respect,  figure  1U 
.shows  a  cross-section  view  of  the  "cartridge"  of  a  typical  crystal-type  reproducer.  _  _ 

An  interesting  reproducer  which  serves  to  illustrate  at  once  the  moving-coil  principle 


13-44  ELECTROMECHANICAL-ACOUSTIC  DEVICES 


CRYSTAL 

ELEMENT 


TORSIONAL 
f  DRIVE  WIRE 


METAL 
BEARINGS 

\ 


CRYSTAL 
MOUNTING  PADS 


BEARING  SUPPORTS 

(RUBBER)  STYLUS 

Pro.  10.    Brush  Development  Co.  PL-20  Crystal  Reproducer 

as  applied  to  both  vertical-  and  lateral-type  reproducers  is  the  Western  Electric  9A  repro- 
ducer, a  phantom  view  of  which  is  shown  in  Fig.  11,  This  is  a  universal  reproducer,  that 
is,  one  which  will  reproduce  either  vertical  or  lateral  recordings  by  changing  the  electrical 


pFia.  11.     Western  Electric  Co.  9A  Universal  Reproducer 

relationship  of  the  two  voltage  generating  coils.    Figure  12  shows  the  response  charac- 
teristic of  this  device  on  both  types  of  recordings. 

Jeweled  styli  have  become  commonplace  in  many  makes  of  reproducers.    Their  contours 


<o-65| 


100         200  SCO         1000      2000          50OO     10,OOO  20,000 

FREQUENCY  IN  CYCLES  PER  SECOND 

PIG."  12.     Response  vs.  Frequency  Characteristic  of  Typical  9A  Repro- 
ducer 


vary,  but  in  general  the 
radius  of  the  spherical 
tip  is  held  to  between 
0.002  and  0.003  in.  The 
larger  tip  radii,  though 
permissible  and  even  de- 
sirable in  certain  types. 
of  lateral  reproducers,  in. 
general  induce  serious. 
distortion  products. 

They  do,  of  course,  re- 
duce wear  on  the  record. 


SOURCES  OF  DISTORTION  13-45 

26.  SOURCES  OF  DISTORTION 

RECORDING.  In  all  recording  instruments  sources  of  distortion  common  to  most 
electromechanical  transducers  are  present.  Magnetic  elements  do  not  always  behave 
linearly,  and  mechanical  elements  are  frequently  free  to  respond  in  modes  other  than  the 
desired  one.  In  general,  however,  distortion  due  to  these  effects  can  be  minimised  by  good 
engineering  and  design.  Recorders  intended  for  use  in  a  variety  of  recording  media  do 
not  always  meet  the  same  impedance  at  the  cutting  point.  Not  only  does  this  impedance 
vary  from  one  medium  to  another;  it  may  vary  also  with  depth  of  cut  in  a  given  medium. 
It  has  been  customary,  therefore,  to  keep  the  driving-point  impedance  sufficiently  high 
so  that  the  recording  medium  has  little  or  no  effect  on  the  recorder  characteristics.  The 
maximum  force  on  the  stylus  in  cutting  is  developed  by  the  reaction  of  the  record  material 
to  be  cut  away.  This  force  is  normal  to  the  motion  of  the  stylus,  and  yielding  by  the  stylus 
arm  in  this  direction  occurs  in  the  neighborhood  of  the  stylus-arm  resonance  frequency. 
This  motion  is  introduced  on  the  time  axis  of  the  recording,  and  intermodulation  with 
other  frequencies  results. 

A  less  important  source  of  distortion  which  intrudes  also  on  the  smoothness  of  cutting 
is  the  design  of  the  stylus.  The  dimensions  of  the  stylus,  like  so  many  other  elements  in 
recording  systems,  are  a  compromise  between  many  factors,  including  linear  velocity  and 
recording  amplitude.  In  order  to  provide  adequate  ruggedness  at  the  tip,  a  short  bevel 
at  a  rather  blunt  angle  is  provided,  as  is  shown  in  Fig.  7.  The  slope  of  this  bevel  marks 
the  maximum  slope  that  can  be  tolerated  in  vertical  recording  without  having  the  heel  of 
the  stylus  abrade  the  modulated  groove.  The  corresponding  critical  angle  in  lateral 
recording  is  approximately  45°,  which  means  that  the  maximum  recorded  vibratory 
velocity  must  never  be  more  than  the  linear  velocity  of  the  record. 

PROCESSING  AND  DUPLICATION.  Although  electrodeposition  is  one  of  the 
most  accurate  methods  known  for  duplicating  a  surface,  several  sources  of  distortion 
present  themselves.  Plating  on  the  face  of  copper  stampers  of  abrasion-resistant  chro- 
mium or  other  metallic  finish  has  a  finite  thickness  sometimes  comparable  with  the  modu- 
lations of  the  grooves.  When  this  process  is  repeated  a  number  of  times,  as  is  common 
commercial  practice,  high  frequencies  are  all  but  obliterated.  Stampers  are  occasionally 
burled  before  or  after  plating.  This,  of  course,  very  rapidly  erases,  in  a  random  manner, 
much  of  the  higher  frequencies. 

REPRODUCING.  The  distortions  introduced  in  recording  and  processing  can  be 
minimized  by  good  design  and  careful  handling.  In  the  interest  of  commercial  expediency 
these  factors  are  sometimes  overlooked.  In  reproducers,  however,  there  are  in  the  present 
state  of  the  art  several  distortion-producing  factors  which  might  be  characterized  as  in- 
herent. The  first  of  these,  termed  "tracing"  distortion,  is  due  to  the  fact  that  the  re- 
producer stylus  has  finite  size.  The  curve  traced  by  the  center  of  a  spherical-tip  stylus  in 
tracing  a  sinusoidally  modulated  groove  is  not  sinusoidal.  This  effect  increases  rapidly 
as  the  minimum  radius  of  curvature  of  the  recorded  wave  approaches  the  radius  of  the 
stylus  tip.  Practical  considerations  prevent  relief  by  the  simple  expedient  of  making  the 
stylus  radius  smaller. 

The  TYnm'rnum  radius  of  curvature  of  a  sine-wave  groove  is  given  by 


where  R  =  niinimurn  radius  of  curvature  (inches),  V  —  linear  speed  of  groove  (feet  per 
minute),  A  =  amplitude  (inches),  and  /  =  frequency  (cycles  per  second).  The  linear 
speed  V  of  a  12-in.  record  at  the  standard  turntable  speed  for  phonographs,  78.26  rpm, 
varies  between  1.3  and  3.8  ft  per  sec.  At  the  standard  broadcast  transcription  speed  of 
33.33  rpm  the  linear  speed  of  a  16-in.  record  varies  between  1.2  and  2.4  ft  per  sec.  At  liigh 
frequencies,  therefore,  for  even  extremely  small  amplitudes  the  radius  of  curvature  of 
the  modulated  groove  may  be  comparable  with  that  of  the  stylus  attempting  to  trace  it. 
Fortunately,  the  distribution  of  energy  in  speech  and  music  spectra  is  such  as  to  alleviate 
this  situation.  However,  hi  the  interest  of  noise  reduction  and  standardization  of  recording 
techniques  it  is  an,  accepted  procedure  to  accentuate  high  frequencies  in  recording  with 
corresponding  attenuations  in  reproduction. 

In  lateral  reproduction  the  problem  is  further  complicated  by  what  has  been  described 
as  "pinch*'  effect;  i.e.,  since  conventional  lateral  reproducers  have  no  vertical  compliance 
the  stylus  "pinches"  between  the  walls  of  the  grooves  when  the  linear  velocity  drops  below 
a  certain  point.  This  phenomenon  results  from  the  fact  that  the  cutting  surface  of  the 
recording  stylus  is  always  perpendicular  to  the  unmodulated  groove.  A  constriction. 
therefore  results  in  the  width  of  those  portions  of  a  modulated  groove  which  are  at  an 
angle  to  the  direction  of  the  unmodulated  groove,  i.e.,  at  all  portions  of  the  modulated 


13-46  ELECTKOMECHANICAL-ACOUSTIC  DEVICES 

grooves  except  the  maxima  and  minima  and  points  of  inflection.  Some  relief  is  afforded 
by  providing  appropriate  vertical  compliance  in  lateral  reproducers,  and  in  such  repro- 
ducers it  is  advantageous  to  use  an  oversize  reproducing  stylus  which  never  rides  on  the 
groove  bottom  but  is  always  positively  driven  by  the  side  walls.  It  has  been  shown  that 
in  this  way  even  harmonics  can  be  eliminated  from  "tracing"  distortion. 

Another  source  of  distortion  in  reproduction  is  due  to  "tracking  error."  Tracking  error 
may  be  defined  as  the  angle  between  the  vibration  axis  of  the  mechanical  system  and  the 
tangent  to  the  groove  being  reproduced.  This  angle  results  from  the  conventional  device 
of  pivoting  the  reproducer  arm  at  a  fixed  point.  The  aforementioned  vibration  axis  can 
therefore  be  truly  tangent  to  the  record  groove  at  only  one  radius.  In  lateral  reproduction 
when  the  tracking  error  is  large  a  sinusoidal  wave  is  not  traced  sinusoidally.  This  effect 
is  minimized  in  lateral  reproduction  by  the  well-known  device  of  an  offset  reproducer 
head  or  by  making  the  pivoted  reproducer  arm  very  long.  Mechanisms  which  keep  the 
reproducer  tangent  at  all  times  are  employed  in  a  few  reproducing  devices,  eliminating 
this  source  of  distortion  entirely.  In  vertical  reproduction,  distortion  due  to  tracking 
error  is  negligible. 

The  characteristics  of  the  medium  of  which  disks  are  made  contribute  to  the  distortion 
in  reproduction.  The  modulated  groove  which  drives  a  reproducer  stylus  has  a  finite 
impedance  which  at  some  frequencies  may  be  comparable  with  or  less  than  the  stylus- 
point  impedance.  With  modern  high-quality  reproducers  the  compliance  of  the  record 
material  resonates  with  the  vibratory  mass  of  the  reproducer,  frequently  producing  a 
peaked  response  at  some  high  frequency,  above  which  the  response  very  rapidly  declines. 
The  most  outstanding  examples  of  the  effect  of  record  characteristics  on  response  are  found 
in  the  instantaneous  types  of  playback  disk.  The  materials  of  which  these  disks  are  made 
have  a  high  compliance.  The  losses  at  high  frequencies,  particularly  at  low  record  veloci- 
ties, are  serious  even  with  the  best  of  commercial  reproducers.  Compensation  for  such 
losses  is  difficult  because  the  magnitude  of  the  loss  varies  widely,  increasing  as  the  linear 
velocity  decreases. 

BIBLIOGRAPHY 

T.  A.  Edison,  U.  S.  Pat.  200,521. 

H.  M.  Stoller,  J.  S.M.P.E.,  1928. 

O.  J.  Zobel,  B.S.T.J.,  1928. 

S.  P.  Mead,  B.S.TJ.,  1928. 

H.  A.  Frederick,  J.  A.S.A.,  1931. 

E.  C.  Wente  and  A.  L.  Thuras,  J.  A.S.A.,  1931. 

W.  C.  Jones  and  L.  W.  Giles,  J.  S.M.P.B.,  1931. 

W.  C.  Jones,  J.  S.M.P.E.,  1931. 

E.  G.  Wente,  Phys.  Rev.,  1922. 
H.  F.  Olsen,  J.  A.S.A.,  1931. 

H,  C.  Harrison  and  P.  B.  Flanders,  B.S.T.J.,  1932. 

H.  D.  Arnold  and  I.  B.  Grandall,  Phys.  Reo.t  1917. 

Tucker  and  Paris,  Phil.  Trans.  Roy.  Soc.,  1921. 

A.  V.  Hippel,  Ann.  d.  Physik,  1925. 

W.  Spaeth,  Zeit.  tech.  Phys.,  1925. 

U.  S.  Patent  1,634,210,  1927. 

Olson  and  Massa,  Applied  Acoustics.    P.  Blakiston's  Son  &  Co.,  Philadelphia. 

J.  P.  Maxfield  and  H.  C.  Harrison,  Trans.  A.I.E.E.,  1926. 

C.  F.  Eyring,  J.  A.S.A.,  1930,  1931;  J.  Soc.  Mot.  Pict.  Eng.,  1930. 

H.  L.  Hanson,  J.  S.M.P.E.,  1930. 

F.  L.  Hunt,  J.  A.S.A.,  1931. 

W.  A.  MacNair,  J.  A.S.A.,  1930;  Proc.  I.R.E.,  1931. 

J.  C.  Steinberg,  J.  A.S.A.,  1929. 

W.  C.  Sabine,  Collected  Papers  on  Acoustics. 

L.  J.  Sivian,  H.  E.  Dunn,  and  S.  D.  White,  J.A.S.A.,  1931. 

J.  P.  Maxfield,  J.  S.M.P.E.,  1930. 

-L.  Cowan,  Recording  Sound  for  Motion  Pictures.    McGraw-Hill,  Chapters  4   16 

E.  O.  Scriven,  J.  S.M.P.E.,  1928;  B.S.T.J.,  1934, 
W.  P.  Button  and  S.  Read,  Jr.,  J.  S.M.P.E.,  1931. 
S.  Read,  Jr.,  J.  S.M.P.E.,  1933. 

R.  A.  Miller  and  H.  Pfannenstiehl,  J".  S.MJP.E.,  1932. 

H.  A.  Frederick,  J.  S.M.P.E.,  1928,  1932. 

P.  Wilson  and  G.  W.  Webb,  Modem  Gramophones  and  Electric  Reproducers.    Cassell  &  Co.  (1929). 

S.  r.  Williams,  J.  Franklin  Inst.,  1926. 

H.  A.  Frederick  and  H.  C.  Harrison,  Trans.  AJ,E.E.r  1933. 

A.  Gizntherschuze,  Zeit.  f.  Phys.,  1926. 
H.  F.  Fnith,  B.S.T.J.,  1932. 

W.  Blum  and  G.  B.  Hogaboom,  Principles  of  Electroplating  and  Electro  forming.    McGraw-Hill. 

F.  C.  Barfeon,  J.  S.M.P.E.V  1934. 

L.  F.  Rahm,  Plastic  Molding,  McGraw-Hill  (1933). 
K  W.  Kellogg,  Trans.  AJ.E.E.,  1927. 
J.  P.  MaxfieM,  U.  S.  Pat.  RE  18,228. 

B.  Nesper,  Nimm  Schallplatten  seiner  auf.    Franckh'sche  Verlagshandlung,  Stuttgart. 
U»  fc>»  .tut*  1,7 54,935  (.1930). 

B.  a  Pat.  1,421,045  (1922). 

AvCL  Kefev  J.  A.S.A.,  VoL  8  (1937). 


PHOTOGRAPHIC   SOUND   RECORDING 


13-47 


A.  C.  Keller,  J.  S.M.P.E.,  1937. 

M.  J.  DiToro,  J.  S.M.P.E.,  Nov.  1937. 

B.  Olney,  Electronics,  November  1937. 

L.  Vieth  and  C.  F.  Wiebusch,  J.  S.M.P.E.,  January  1938. 

J.  A.  Pierce  and  F.  L.  Hunt,  J.  S.M.P.E.,  August  1938. 

H.  A.  Henning,  Bett  Lab.  Rec.>  October  1940. 

W.  D,  Lewis  and  F.  L.  Hunt,  J.  A.S.A.,  January  1941. 

L.  Fleming,  J.  A.S.A.,  January  1941. 

B.  B.  Bauer,  Electronics,  March  1945. 

B.  B.  Bauer,  J.  A.S.A.,  April  1945. 

H.  E.  Boys,  J.  S.M.P.E.,  June  1945. 

W.  S.  Bachman,  Electronic  Industries,  July  1945. 


PHOTOGRAPHIC  SOUND  RECORDING 

By  C.  R,  Keith 

The  photographic  method  of  sound  recording  finds  its  greatest  field  in  sound  pictures, 
the  sound  record  being  usually  on  the  same  film  as  the  picture  or  at  times  on  a  separate 
film  operating  synchron- 
ously with  the  picture  film. 
Film  records  are  particu- 
larly adapted  to  sound  pic- 
tures since  these  films  are 
usually  made  from  numer- 
ous short  "takes"  which  are 
edited  and  then  spliced  to- 
gether. The  sound  and 
picture  are  usually  photo- 
graphed on  separate  films, 
except  for  newsreel  record- 
ing where  the  sound  and 
picture  are  commonly  taken 
on  the  same  film.  Usually 
music  and  sound  effects  are 
recorded  separately  from 
the  dialog;  the  various 
sound  tracks  are  then  re- 
recorded in  synchronism 
with  the  finally  cut  picture 
film,  to  make  a  sound  nega- 
tive which  is  used  for  makT 
ing  release  prints. 

Two  types  of  sound-on- 
film  records  are  in  common 
use  today.  One  is  the  vari- 
able-density type — a  series 
of  striated  bands  as  shown 
in  Fig.  1  (a)-(d).  The  other 
is  the  variable-area  type 
shown  in  Fig.  1  (e)-(i) — a 
serrated  band  with  its 
toothlike  projections.  Both 
place  the  record  on  a  narrow 
strip  of  the  film  at  one  side 
of  the  picture  as  in  the 
sketch  of  a  composite  print 
with  a  variable-density 
sound  track  (Fig.  2).  As 
the  sound  track  must  be 
played  at  uniform  speed 
while  the  picture  progresses 
with  intermittent  motion, 
it  is  displaced  forward  along 
the  film  15  in.  from  the 


FIG.  1.    Types  of  Sound  Tracks 


corresponding  picture,  so  that  the  momentary  difference  in  film  velocity  can  be  taken  up 


13-48  ELECTROMECHANICAL-ACOUSTIC  DEVICES 


-t-0.100  In. 


by  a  free  loop.    Both  types  of  sound  track  can  be  reproduced  in  the  same  machine  with- 
out any  change  in  the  machine  or  in  the  electrical  equipment. 

The  most  commonly  used  sound  tracks  are  illustrated  in  Fig.  1.  Films  exhibited  in 
theaters  ordinarily  have  single  100-mil  tracks  such  as  (a),  (b),  (e),  (/),  or  (g).  Original 
sound  records  are  often  made  with  100-mil  pushpull  tracks 
such  as  (c)  or  (i)  or  more  commonly,  when  the  sound  is 
recorded  on  a  film  separate  from  the  picture  film,  with  a 
200-mil  pushpull  track  such  as  (d)  or  (h) . 

There  are  many  methods  of  producing  such  film  sound 
tracks:  the  light- valve  method  (Western  Electric),  the  reflect- 
ing-galvanometer  method  (RCA),  the  flashing-lamp  method 
(Fox-Case),  the  Kerr  cell  (Klangfilm\  and  many  variations  of 
these  methods.  The  flashing  lamp  and  the  Kerr  cell  are  used 
only  for  variable-density  tracks,  but  the  light  valve  and  the 
reflecting  galvanometer  may  be  used  for  either  variable-density 
or  variable-area  tracks. 

27.  LIGHT-VALVE  RECORDING  SYSTEM 

In  the  light-valve  system  the  light  from  an  incandescent 
source  is  made  to  fall  on  a  light  valve,  Fig.  3,  formed  of  two 
strips  of  Duralumin  0.0005  in.  thick  by  0.006  in.  wide  spaced 
0.001  in.  apart.  These  are  placed  in  a  magnetic  field  and  carry 
the  speech  currents.  Since  the  two  ribbons  carry  current  in 
opposite  directions  they  move  together  or  apart  as  the  current 
varies.  In  commercial  practice  the  ribbons  are  tuned  or  reso- 
nated to  about  9500  CRS.  The  response  in  the  frequency  range 
near  resonance  depends  on  the  damping,  and  in  early  types  of 
light  valves  the  response  at  resonance  was  20  to  25  db  higher  than  at  low  frequencies. 
This  variation  in  response  can  be  reduced  to  1  or  2  db  by  using  a  feedback  circuit  in  which 
signal  voltage  across  the  light-valve  ribbons  is  amplified  and  applied  to  the  valve  input 
in  opposite  phase.  However,  the  more  recent  light  valves  have  sufficiently  high  flux  den- 
sity (30,000  gausses)  so  that  the  resonance  peak  is  only  about  6  db.  Additional  damp- 
ing is  obtained  by  passive  networks  in  the  valve  input  circuit,  so  that  the  resonance  peak 
may  be  reduced  to  about  1  db.  These  measures  also  almost  entirely  eliminate  extraneous 
vibrations  of  the  light-valve  ribbons  when  excited  by  steep  transients. 

In  early  types  of  light  valves  both  ribbons  were  in  the  same  plane  so  that  it  was  possible 
for  them  to  strike  each  other  when  caused  to  vibrate  at  large  amplitudes,  thereby  increasing 
the  normal  overload  distortion.  This  difficulty  is  avoided  in  later  types  by  mounting  the 
two  ribbons  in  parallel  planes  separated  by  about  the  thickness  of  one  ribbon.  In  addition 
to  preventing  distortion  due  to  ribbons  striking  each  other,  this  "biplanar"  construction  re- 
duces the  possibility  of  damage  to  ribbons  through  overmodulation. 

When  the  light-valve  ribbons  are  focused  directly  on  the  moving  film  the  exposure  at 
high  frequencies  does  not  correspond  to  the  motion  of  the  ribbons  if  the  velocity  of  either 
edge  of  the  image  is  comparable  to  that  of  the  film.  The  exposure  given  to  the  film  is 


H  Scanning 
-H  H-Beam 

0.084  In. 

FIG.    2.      Position  of  Sound 
Track  on  35-mm  Film 


Lamp 


Ught  Vate 
FIG.  3.     Optical  Schematic  of  Typical  Light-valve  Recording  System 


Objective 
Lens 


elm 


determined  by  the  time  required  for  any  point  on  the  film  to  pass  through  the  image  of 
the  light-valve  slit;  in  other  words,  the  exposure  (to  a  first  approximation)  is  the  product 
of  time  and  intensity.  If  the  frequency  being  recorded  is  low,  so  that  the  velocity  of  the 
ribbons  is  small  compared  with  that  of  the  film,  the  variations  in  film  exposure  will  accu- 
rately correspond  to  the  light-valve  modulation.  As  the  frequency  becomes  higher,  how- 
ever, tke  velocity  of  the  ribbons  increases,  being  substantially  proportional  to  frequency 


LIGHT-VALVE  RECORDING  SYSTEM 


13-49 


for  constant  electrical  power  input  to  the  light  valve,  until  at  the  highest  audio  frequencies 
the  ribbon- velocity  effect  may  become  important.  For  a  sine-wave  signal  just  sufficient 
to  fully  modulate  a  two-ribbon  valve  spaced  to  0.001  in.,  the  peak  ribbon  velocity  becomes 
equal  to  the  film  velocity  at  about  6000  cycles.  This  results  in  a  loss  of  high-frequency 
response  which  also  varies  with  the  average  ribbon  spacing.  Consequently  when  a  high- 
frequency  wave  is  superposed  on  a  low-frequency  wave,  the  high-frequency  response 
varies  during  each  cycle  of  the  low  frequency. 

Distortion  of  this  type  is  not  indicated  by  harmonic  measurements  since  it  occurs 
mainly  at  frequencies  whose  harmonics  would  be  greatly  attenuated  by  other  elements 
in  the  system,  such  as  the  width  of  the  reproducing  slit.  Such  distortion  can,  however, 
be  detected  by  "intermodulation"  tests  in  which  two  frequencies  are  recorded  simultan- 
eously (one  high  and  one  low)  and  the  variation  in  high-frequency  response  measured. 
In  usual  practice  the  amplitude  of  the  low  frequency  (60  cycles)  is  four  times  that  of  the 
high  frequency  (7000  cycles) .  The  reproduced  wave  is  passed  through  a  high-pass  filter, 
eliminating  the  60-cycle  component.  When  this  wave  is  rectified  a  new  set  of  low  fre- 
quencies (60,  120,  180.  . .  cycles)  is  produced,  owing  to  the  presence  of  distortion  frequen- 
cies in  the  high-frequency  wave.  The  average  amplitude  of  this  new  low-frequency  wave 
is  a  convenient  measure  of  the  distortion.  In  a  suitably  designed  light- valve  modulator 
the  intermodulation  distortion  may  be  reduced  to  4  per  cent  or  less,  corresponding  to 
approximately  1  per  cent  harmonic  distortion  as  measured  in  a  system  in  which  distortion 
and  transmission  do  not  vary  with  frequency.  This  is  accomplished  by  reducing  the 
effective  image  width  to  0.00025  in.  or  less,  usually  by  means  of  a  short-focus  cylindrical 
lens  placed  close  to  the  film  surface. 

NOISE-REDUCTION  SYSTEM.  A  noise-reduction  system  is  commonly  used  in  film 
recording  in  order  to  lower  the  level  of  background  noise  when  it  is  not  masked  by  rela- 
tively loud  sounds.  This  system  is  based  on  the  fact  that  in  a  variable-density  sound 
track  the  background  noise  output  from  a  light  print  is  greater  than  that  from  a  dark 
print  when  reproduced  with  the  same  gain  setting  of  the  amplifier.  However,  when  a 
print  is  made  darker  by  merely  increasing  the  exposure  in  the  printer,  both  the  ground 
noise  and  the  wanted  sound  are  reduced  in  approximately  the  same  ratio  so  that  no  im- 
provement in  the  signal-to-noise  ratio  results.  The  desired  result  is  obtained  by  reducing 
the  average  exposure  of  the  negative  during  periods  of  low  modulation  without  reducing 
the  amount  of  light  modulated  by  the  signal. 

In  the  Western  Electric  system  of  noiseless  recording  the  mean  spacing  of  the  light- 
valve  ribbons  is  made  to  vary  so  that  as  modulation  is  impressed  on  the  valve  the  mean 
spacing  increases  sufficiently  to  accommodate  the  increasing  input.  The  spacing  between 
light-valve  strings  is  mechanically  adjusted  to  0.001  in.,  and  a  source  of  direct  current  is 
connected  to  the  strings  sufficient  to  reduce  the  spacing  to  0.0003  in.  (for  10-db  noise 
reduction) .  In  addition  to  this  fixed  bias  a  varying  bias  proportional  to  the  envelope  of 
the  signal  wave  increases  the  ribbon  spacing  as  the  signal  current  increases.  The  lower 
average  exposure  of  the  negative  during  periods  of  low  signal  levels  produces  a  positive 
with  relatively  high  average  density  during  such  periods.  The  ground  noise  is  thereby 
reduced  when  the  signal  is  low,  but,  since  the  valve  modulation  due  to  the  signal  is  un- 
changed, the  effective  signal-to-noise  ratio  is  increased. 


Main 
Ampli- 
fier 


£ S. 

l-l  [f 


Ampli- 
fier 


Recti- 
fier 


Timing 
RIter 


Modu- 
lator 


Ampli- 
fier 


Recti- 
fier 


Biter 


FIG.  4.     Block  Schematic  of  Noise-reduction  System 


A  block  schematic  of  a  typical  noise-reduction  circuit  is  shown  in  Fig.  4.  The  timing 
filter  not  only  serves  to  remove  audible  frequencies  from  the  rectified  signal  but  also  shapes 
the  bias  wave  so  that,  for  rapidly  fluctuating  signals  such  as  speech,  the  valve  spacing 
increases  at  about  the  same  rate  as  the  signal  increases,  but  the  spacing  decreases  at  a 
considerably  slower  rate.  The  oscillator,  modulator,  second  amplifier,  and  rectifier  merely 
form  a  convenient  means  of  amplifying  the  fixed  and  variable  bias  currents. 


13-50  ELECTROMECHANICAL-ACOUSTIC  DEVICES 


28.  REFLECTING-GALVANOMETER  RECORDING  SYSTEM 

The  refiecting-galvanometer  method,  commonly  used  for  producing'variable-area  sound 
tracks,  utilizes  a  recording  device  that  operates  on  the  principle  of  the  mirror  oscillograph. 
Light  is  reflected  from  the  oscillating  mirror  of  the  recorder  and  is  passed  through  a 
narrow  slit  onto  the  film.  The  resulting  sound  track  has  constant  density  but  variable 


(«)  CO 

FIG.  5.    (a)  Optical  Schematic  of  Typical  Galvanometer  Recording  System,    (b)  Recording  Gal 
eter.    (c)  Armature  Damping  in  Recording  Galvanometer,     (d)  (e)  (/)  Typical  Mask  and  Shu1 


rangements  Used  in  Galvanometer  Recording  System. 


ivanom- 
itter  Ar- 


width.  A  widely  used  type  of  light  modulator  accomplishes  this  by  the  use  of  a  triangular 
beam  of  light  which  is  caused  to  move  at  right  angles  to  the  axis  of  the  slit  by  the  recording 
galvanometer,  so  that,  as  it  vibrates,  the  length  of  the  illnTnj.Ti.ed  portion  of  the  slit  varies. 
The  RCA  variable-width  light  modulator,  Fig.  5 (a),  consists  essentially  of  an  incan- 
descent lamp,  a  system  of  lenses  to  direct  the  light,  an  aperture  and  slit  to  limit  the  light, 


FLASHING-LAMP,   KERR  CELL  RECORDING  SYSTEMS      13-51 

and  a  reflecting-mirror  galvanometer  to  modulate  the  light.  An  image  of  filament  A  is 
formed  at  the  galvanometer  mirror  F  by  the  combination  of  lenses  B  and  E.  The  aper- 
ture C,  shown  as  the  shaded  area  in  Fig.  5(d)T  limits  the  light  projected  to  the  galvanometer 
mirror.  Lens  E  forms  an  image  of  aperture  <?  on  slit  H.  Vibration  of  the  galvanometer 
mirror  moves  this  light  beam  back  and  forth  across  the  slit  so  that  the  length  of  the  illum- 
inated portion  of  the  slit  is  proportional  to  the  angular  deflection  of  the  mirror.  A  reduced 
image  of  the  slit  is  formed  on  the  film  K  by  objective  lens  J. 

^  Ground  noise  is  reduced  in  this  modulator  by  cutting  off  the  light  from  the  ends  of  the 
slit  during  periods  of  low  modulation.  The  two  magnetically  operated  shutter  vanes  D 
are  used  for  this  purpose,  producing  the  varying  black  margins  shown,  for  example,  in 
Fig.  l(g).  Current  for  operating  the  noise-reduction  shutters  is  derived  from  the  signal 
and  is  proportional  to  the  envelope  of  the  signal  wave.  During  periods  of  no  modulation 
the  clear  area  is  reduced  to  a  width  of  about  0.002  in. 

Other  common  accessories  in  this  modulator  are  a  visual  monitor  system  and  an  ultra- 
violet filter.  The  visual  monitor  is  provided  by  forming  an  image  of  one  edge  of  the  aper- 
ture C  on  a  monitor  card.  Vibration  of  the  galvanometer  mirror  causes  the  image  to 
lengthen  in  proportion  to  the  mirror  deflection.  By  means  of  suitably  spaced  lines  on  the 
card  the  operator  is  enabled  to  judge  when  the  galvanometer  deflection  reaches  the 
maximum  allowable  without  exceeding  the  sound-track  width  (0.076  in.)  that  can  be 
scanned  by  the  slit  in  the  reproducing  machine.  The  ultraviolet  filter  I  restricts  the  film 
exposure  to  a  narrow  band  of  wavelengths  in  the  neighborhood  of  3650  A,  thereby  reducing 
image  spread  due  to  scattering  of  light  in  the  film  emulsion.  Recent  fine-grain  films  have 
also  contributed  to  improvement  in  image  quality. 

The  recording  galvanometer,  Fig.  5(6)r  consists  of  a  pair  of  permanent  magnets,  pole 
pieces,  balanced  armature  A,  signal  and  bias  coils  C,  mirror  support,  and  mirror  Af.  A 
pair  of  phosphor-bronze  springs  hold  a  groove  ia  the  ribbon  support  against  a  knife  edge 
on  the  end  of  the  armature.  Current  through  either  the  signal  or  biasing  coils  polarizes 
the  armature,  causing  it  to  be  attracted  to  the  pole  piece  of  opposite  polarity  and  by  its 
lateral  motion  rotating  the  mirror  support  and  mirror. 

One  of  the  serious  problems  in  electromechanical  apparatus  is  the  provision  of  suitable 
damping.  Oil  damping,  although  widely  used  in  oscillograph  galvanometers,  has  a 
number  of  objections,  among  which  are  the  change  of  viscosity  with  temperature,  relatively 
large  mass  required  for  the  damping  obtained,  and  difficulty  of  avoiding  leakage.  The 
damping  in  this  galvanometer  is  obtained  by  utilizing  the  properties  of  tungsten-loaded 
rubber  so  mounted  that  damping  is  obtained  in  the  frequency  range  near  armature  reso- 
nance, but  without  affecting  the  low-frequency  response  of  the  galvanometer.  As  shown, 
in  Fig.  5(c),  two  small  pieces  of  tungsten-loaded  rubber  R  are  cemented  to  the  sides  of 
the  armature  A,  and  a  bronze  yoke  presses  against  their  outer  faces.  At  low  frequencies 
the  yoke  and  pads  move  with  the  armature,  but  at  high  frequencies  the  inertia  of  the  yoke 
causes  it  to  tend  to  stand  still  while  the  armature  vibrates  inside  it,  compressing  the  rubber 
and  damping  the  peak. 

Other  types  of  sound  tracks  can.  be  made  with  the  same  modulator  by  means  of  masks 
of  other  shapes.  For  example  a  class  A  pushpull  variable-width  tra^k  may  be  made  with 
the  mask  in  Fig.  5  (e) .  Class  B  pushpull  variable-width  tracks  may  be  made  with  the  mask 
shown  in  Fig.  5(/).  Variable-density  tracks  can  also  be  made  by  a  modificatJoa  of  this 
modulator  in  which  the  uniformly  illuminated  triangles  are  replaced  by  a  peoumbra 
designed  to  give  a  linear  gradation  of  light  intensity.  The  galvanometer  mirror  causes 
this  penumbra  to  move  across  the  fixed  slit,  varying  the  light  transmitted  to  the  film  in 
proportion  to  the  mirror  deflection. 

29.  FLASHING-LAMP  AND  KERR  CELL  RECORDING  SYSTEMS 

Although  not  widely  used  at  the  present  time,  the  flashing-lamp  method  of  recording 
has  the  advantage  of  extreme  simplicity-  The  gaseous  discharge  is  concentrated  in.  a 
relatively  small  area  and  so  designed  that  the  current  is  proportional  to  the  signal  voltage. 
An  early  type  of  flashing  lamp,  called  Aeolight,  is  a  two-element  tube  containing  an  inert 
gas  such  as  helium.  One  of  the  elements  is  of  nickel;  the  other  is  coated  with  barium  and 
strontium  oxides.  When,  sufficient  voltage  is  applied  1,0  the  electrodes,  ioniEation  takes 
place  and  a  concentrated  glow  appears  at  the  cathode.  The  intensity  of  tfee  light  so  pro- 
duced increases  in  proportion  to  the  increase  in  applied  voltage.  la  operation,  sufficient 
polarizing  voltage  is  applied  to  the  tube  to  give  the  required  average  exposure  of  ^  the 
negative,  and  sound  voltages  are  superposed  on  this  polarizing  voltage.  Since  the  light 
output  is  comparatively  low,  the  lamp  is  usually  arranged  for  direct  iHiimination  of  the 
£lm  rather  than  by  means  of  the  usual  lens  system.  A  fixed  slit  0.0008  in.  by  0,100  in., 


13-52  ELECTKOMECHANICAL-ACOTJSTIC  DEVICES 

made  by  engraving  the  silvered  surface  of  a  small  quartz  block,  is  located  within  less  than 
0.001  in.  of  the  film.  The  low  light  intensity  obtained  from  flashing-lamp  light  modulators 
results  in  a  very  low  value  of  negative  exposure  and  is  in  the  region  known  as  the  "toe" 
of  the  H  and  D  curve.  Prints  made  from  these  negatives  must  be  printed  on  the  "toe" 
of  the  positive  H  and  D  curve  in  order  to  minimize  distortion.  Consequently  the  per- 
missible modulation  ratio  is  reduced  and  the  volume  range  is  less  than  for  the  fully  exposed 
type  of  record. 

Another  relatively  low-intensity  light  modulator  utilizes  the  Kerr  electro-optical  effect. 
A  glass  cell  containing  nitrobenzol  is  placed  between  two  Nicol  prisms,  and  an  electrostatic 
field  is  applied  to  electrodes  on  either  side  of  the  cell.  Light  from  an  incandescent  lamp  is 
plane  polarized  by  the  first  Nicol  prisrn,  and  the  second  prism  is  so  arranged  that,  when  no 
polarizing  voltage  is  applied  to  the  cell  electrodes,  no  light  is  transmitted.  Since  the  plane 
of  polarization  of  light  transmitted  through  the  cell  is  rotated  as  the  polarizing  voltage  is 
increased,  the  transmitted  light  also  is  increased.  However,  the  relation  between  trans- 
mitted light  and  applied  voltage  is  not  linear  and  introduces  appreciable  distortion  at 
high  modulation. 

30.  SOUND-ON-FILM  REPRODUCING  SYSTEMS 


In  the  reproduction  of  either  of  the  two  normal  types  of  single  sound  tracks,  i.e.,  var- 
iable-density or  variable-area,  the  light  from  an  incandescent  filament  is  made  to  fall  on  a 

fixed  slit  which  is  imaged  on 
Slit  the  ^  as  a  line  of 


Condenser 
Lens 


lUght 
Source 


Film/ 
Plane 


)  The  "motion  picture"  type  of  optical  system 


Condenser 


Objective  Lens 


Film./ 
Plane 


The  "stereoptlcon*1  type  of  optical  system 
FIG.  6.     Typical  Film  Reproducer  Optical  Systems 


common  optical  systems  are 
shown  in  Fig.  6.  In  either, 
the  light  passing  through  the 
film  falls  directly  on  a  photo- 
electric cell  or  is  transmitted 
to  it  through  a  suitable  com- 
bination of  lenses  and  prisms. 
Usually  the  photocell  optical 
system  subtends  a  solid  angle 
of  less  than  2-rr  radians  as 
measured  from  a  point  on  the 
sound  track.  Light  passing 
through  the  film  emulsion  is 
scattered  through  a  variable 
solid  angle  depending  on  the 
density  (Callier  effect),  so 


that  at  the  higher  densities  a  smaller  proportion  of  the  transmitted  light  reaches  the 
photocell.  This  gives  the  effect  of  increased  density  gradation  (gamma)  and  must  be 
taken  into  account  in  processing  variable-density  sound  tracks. 

Pushpull  sound  tracks  require  two  photocells  and  suitable  means  for  reversing  the  phase 
of  one  track  with  respect  to  the  other.  A  typical  pushpull  reproducer,  simplified,  is  shown 
in  Fig.  7. 

Double  Cathode 

Photocell^  Push  PuH 

Flint  J         Transformer' 


Condenser 


FIG.  7. 


Objective  Lens 

Separation 
Optics 

Push-pull  Reproducer  Optical  System 


In  the  reproduction  of  photographic  sound  tracks  there  are  unavoidable  losses  in  output 
related  to  the  speed  with  which  the  sound  track  is  moved,  and  the  dimensions  and  relative 
positions  of  recording  and  reproducing  slits.  Both  the  recording  and  the  reproducing 
light  beams  should  be  exactly  perpendicular  to  the  direction  of  the  film  motion.  Any 
deviation  from  this  direction  is  called  an  error  in  azimuth  and  introduces  a  loss  at  high 
frequencies.  Another  important  loss  is  due  to  the  impossibility  of  producing  on  the  sound 
track  a  line  of  light  which  is  infinitely  narrow.  These  conditions  produce  loss  of  efficiency 


SOUND-ON-FILM  REPRODUCING  SYSTEMS 


13-53 


at  higher  frequencies  and  in  variable-area  tracks  may  also  introduce  distortion.    Figure  S 
shows  typical  loss  curves  of  these  effects  plotted  from  the  following  equations. 

VARIABLE-DENSITY  SOUND  TRACKS.    Effect  of  image  width  for  zero  azimuth: 


Scanning  loss  in  decibels  -  20  logio 


CD 


where  /  =  frequency  in  cycles  per  second,  0  =  half  imag?  width,  and  V  =  film  velocity. 
Effect  of  azimuth  for  fixed  image  width: 


Scanning  loss  in  decibels  = 


2r/j3  sec  a 


2yfl  tan  a 
V 


(2) 


where  I  =  half  width  of  sound  track,  and  a.  —  azimuth  deviation  angle. 

If  the  azimuth  is  zero  the  last  equation  reduces  to  the  equation  preceding  it.  For  very 
small  angles  (0°  to  6°), 
which  covers  the  cases 
of  practical  interest, 
£  sec  a.  —  I  tan  a.  Hence 
in  the  azimuth-loss  equa- 
tion the  image-width  fac- 
tor becomes  equal  to  the 
azimuth  factor  when  the 
azimuth  deviation  is  equal 
to  the  effective  slit  image 
width;  i.e.,  the  scanning 
loss  for  a  particular  azi- 
muth deviation  is  equal  to 
that  of  an  image  width  of 
the  same  distance  along 
the  length  of  the  film. 


25 
1000 


2  345 

Frequency  in  Cycles  per  Second 


789  10,000 


FIG.  8.    Loss  Dae  to  Finite  Sfit  Width 


VARIABLE-AREA  SOUND  TRACKS.  For  variable-area  sound  tracks  the  calcula- 
tion of  the  effect  of  the  aperture  on  fundamental  response  and  generation  of  harmonics 
becomes  considerably  more  complicated  and  involves  a  number  of  assumptions  that  are 
not  always  realized  in  practice.  However,  the  general  effects  may  be  obtained  by  assuming 
an  ideal  record  in  which  the  exposed  portion  has  uniform  density  such  as  would  be  realized 
with  an  infinitely  narrow  recording  slit  and  an  ideal  film  emulsion.  The  effect  of  finite 
reproducing-slit  width  (with  no  azimuth  error)  is  a  reduction  in  response  at  high  fre- 
quencies exactly  the  same  as  shown  above  for  variable-density  sound  tracks.  On  the 
other  hand,  when  the  same  ideal  variable-area  record  is  reproduced  by  means  of  a  finite 
slit  not  perpendicular  to  the  direction  of  motion,  the  loss  in  fundamental  amplitude  for  a 
bilateral  track  is  given  by  ' 


-  —    I 
2am         € 

and  the  ratio  of  second  harmonic  to  fundamental  by 


(4) 


where  2e  =  width  of  reproducing  image,  a  —  width  of  unmodulated  unbiased  track,  and 
m  =  per  cent  modulation,  and 


.  (=)•«£) 


(46) 

and  p  =  cotangent  of  angle  of  azimuth  deviation;  also  Ji  and  J2  are  Bessel  functions  of 
the  first  and  second  order  respectively. 

Third  and  higher  harmonics  are  also  produced  by  an  azimuth  deviation  of  the  reproduc- 
ing slit.  Distortion  may  also  be  caused  by  the  finite  width  of  the  recording  slit,  but  it 
may  be  compensated  to  some  extent  by  choosing  a  suitable  density  for  the  print.  The 


13-54  ELECTROMECHANICAL-ACOUSTIC  DEVICES 


2.4 

/ 

^ 

^ 

a 

0 

8 

/ 

/ 

g  1.6 

V 

/ 

/ 

to 

/ 

/\ 

to, 

/ 

/ 

"\ 

Tz 

n  a  — 

ST 

Q  °-4 

0 

^~~* 

<^S 

! 

\ 

)        0. 

4        0 

8      1. 

2       1 

6      2, 

0      2. 

4       2 

8       3. 

2       3. 

6       4. 

photographic  image  spread  in  the  positive,  then,  to  a  first  approximation,  compensates 
for  both  the  finite  recording  slit  width  and  the  negative  image  spread. 

Distortion  may  also  occur  in  the  reproduction  of  variable-area  sound  records  as  the 
result  of  uneven  illumination  along  the  length  of  the  reproducing  slit.  Although  the 
amount  of  such  distortion  depends  on  the  type  of  variable-area  track  and  on  the  type 
and  amount  of  unevenness  of  illumination,  proper  design  of  the  reproducer  optical  system 
and  a  reasonable  degree  of  adjustment  should  result  in  harmonics  not  more  than  3  per 
cent  of  the  fundamental  (for  full  modulation)  and  in  most  cases  considerably  less. 

In  addition,  variable-area  records  will  obviously  be  distorted  if  the  center  line  of  the 
record  in  the  reproducer  is  displaced  from  the  center  line  of  the  slit  image  so  that  part  of 
the  record  is  not  scanned.  It  is  for  this  reason  that  0.076  in.  is  considered  the  maximum 
track  width  for  distortionless  reproduction  since  with  a  standard  0.084-in.  scanning  image 
a  tolerance  of  ±0.004  in.  is  then  allowed  for  film  weave. 

PHOTOGRAPHIC  REQUIREMENTS.  The  principal  photographic  requirement  of 
sound  recording  is  that  the  variation  of  light  from  the  average  amount  transmitted  by  the 

film  in  the  reproducer  must 
be  proportional  to  the  cor- 
responding variation  from 
the  average  amount  of  light 
transmitted  by  the  record- 
ing modulator.  This  gen- 
eral relation  is  true  for 
both  variable-density  and 
variable-area  tracks. 

For  a  variable-density 
track,  the  exposure  varies 
from  point  to  point  along 
the  length  of  the  track. 
Figure  9  shows  the  manner 
in  which  exposure  and  den- 
sity are  related  in  a  typical 
film  emulsion.  The  exact 
shape,  and  particularly  the 
slope  of  the  central  portion 
of  the  H  and  D  curve,  depend  on  the  development  as  well  as  the  photographic  properties 
of  the  film,  but  most  H  and  D  curves  have  a  portion  that  is  essentially  straight  when 
plotted  as  shown.  The  slope  of  the  straight  portion  is  called  gamma,  -y.  If,  when  a  nega- 
tive is  printed,  a  similar  curve  is  drawn  relating  the  exposure  of  the  negative  to  the  density 
of  the  positive,  the  slope  of  the  straight  portion  of  this  curve  is  the  overall  gamma.  It  can 
be  shown  that  for  minimum  distortion  in  variable-density  recording  the  overall  gamma 
should  be  unity.  In  this  computation  factors  such  as  the  Callier  effect  (projection  factor) 
and  the  departure  of  actual  gammas  from  the  values  measured  by  a  sensitometer  must 
be  taken  into  account.  Since  the  positive  film  carries  both  sound  and  picture,  both  images 
must  be  given  the  same  development.  The  positive  gamma  is  chosen  to  be  suitable  for 
the  picture  and  is  usually  about  2.2.  Consequently,  the  negative  of  a  variable-density 
sound  track  is  developed  to  a  gamma  of  about  0.4  when  measured  on  type  2B  sensitometer. 
Although  optimum  values  of  gamma  and  density  for  variable-density  records  can  be 
obtained  with  a  fair  degree  of  accuracy  from  sensitometric  measurements,  the  simplest 
and  most  reliable  determination  of  these  constants  is  made  by  means  of  intermodulation 
tests.  Such  tests  measure  the  non-linear  distortion  in  a  print  made  from  a  negative  on 
which  two  frequencies  (usually  60  and  1000  cycles)  are  simultaneously  recorded.  Not 
only  is  this  form  of  test  more  sensitive  than  a  measurement  of  harmonics  of  a  single 
frequency  but  also  it  corresponds  more  closely  to  audible  distortion  in  commercial 
records. 

In  processing  variable-area  records  the  aim  is  to  obtain  minimum  density  in  the  clear 
area  and  maximum  density  in  the  exposed  area.  These  are  obtained  by  developing  both 
negative  and  positive  to  relatively  high  gammas,  although  excessive  development  may 
result  in  fog  in  the  clear  area. 

Variable-width  records  are  also  subject  to  non-linear  distortion  if  they  are  not  given 
proper  exposure  and  development.  This  distortion  is  frequently  due  to  spreading  of  the 
image  and  produces  an  effect  similar  to  rectification.  Optimum  processing  is  determined 
by  "cross-modulation"  tests  in  which  a  modulated  high  frequency  is  recorded.  Distortion 
is  measured  by  the  amount  of  low  frequency  produced  by  the  photographic  rectification 
of  the  modulated  high  frequency.  Distortion  may  also  be  caused  by  non-linearity  of  the 
modulator  and  may  be  measured  by  harmonic  or  intermodulation  tests. 


Log  Exposure 
FIG.  9.     H  and  D  Curve 


DEFINITION  OF  EFFECTS  13-55 

BIBLIOGRAPHY 

All  references  are  to  the  Journal  of  the  Society  of  Motion  Picture  Engineers  unless  otherwise  noted. 

General  References 

Academy;  of  Motion  Picture  Arts  and  Sciences,  Motion  Picture  Sound  Engineering.    Van  Noetraad. 

New  York  (1938).  ^^ 

Academy  of  Motion  Picture  Arts  and  Sciences,  Recording  Sound  for  Motion  Pictures.    McGraw-Hill, 

New  York  and  London  (1931). 
Talbot,  R.  EL,  Some  Relationships  between  the  Physical  Properties  and  the  Behavior  of  Motion 

Picture  Film,  Vol.  XLV,  3,  p.  209  (September  1945). 

Kellogg,  E.  W.,  The  ABC  of  Photographic  Sound  Recording,  Vol.  XLIV,  3,  p.  151  (March  1945). 
Iroye,  D.  P.,  and  K.  F.  Morgan,  Sound  Picture  Recording  and  Reproducing  Characteristics,  Vol.  XXXII, 

6,  p.  631  (June  1939),  and  Vol.  XXXIII,  1,  p.  107  (July  1939). 
Honan,  E.  M.,  and  C.  R.  Keith,  Recent  Developments  in  Sound  Tracks,  Vol.  XLI,  2,  p,  127  (August 

Light-valve  Method 

MacKenzie,  D.,  Sound  Recording  with  the  Light  Valve,  Trans.  Soc.  Mat.  Pict.  Engrs.,  VoL  XII,  35, 

p.  730  (September  1938). 
Shea,  T.  E.,  W,  Herriott,  and  W.  R.  Goehner,  The  Principles  of  the  Light  Valve,  VoL  XVIII,  6,  p.  697 

(June  1932). 
Frayne,  J.  G.,  and  H.  C.  Silent,  Push-pull  Recording  with  the  Light  Valve,  VoL  XXXI,  1,  p.  46 

(July  1938). 

Refiecting-galvanometer  Method 

Dimmick,  G.  LM  Galvanometers  for  Variable  Area  Recording,  Vol.  XV,  4,  p.  428  (October  1930). 
Batsel,  M.  C.,  and  E.  W.  Kellogg,  The  RCA  Sound  Recording  System,  VoL  XXVIII,  5f  p.  507  (May 

1937). 
Dimmick,  G.  L.,  The  RCA  Recording  System  and  Its  Adaptation  to  Various  Types  of  Sound  Track, 

VoL  XXXIX,  3,  p.  258  (September  1937). 

Kerr  Cell  Method 

Zworykin,  V.,  L.  B.  Lynn,  and  C.  R.  Hanna,  Kerr  Cell  Method  of  Sound  Recording,  Tram.  Soc.  Moi. 
Pict.  Eng.,  VoL  XII,  35,  p.  748  (September  1928). 

Ground-noise  Reduction 

KreuzerT  B.,  Noise  Reduction  with  Variable  Area  Recording,  Vol.  XVI,  6,  p.  671  (June  1031). 

Silent,  H.  C.,  and  J.  G.  Frayne,  Western  Electric  Noiseless  Recording,  VoL  XVIII,  5,  p.  551  {May  1932). 

Kellogg,  E.  W.,  Ground  Noise  Reduction  System,  VoL  XXXVI,  2,  p.  137  (February  1941). 

Scoville,  R.  R.,  and  W.  L.  Bell,  Design  and  Use  of  Noise  Reduction  Systems,  Vol.  XXXVIII,  2,  p. 
125  (February  1942). 

Reproducing  Systems 

Cook,  E.  D.,  The  Aperture  Effect,  VoL  XIV,  6,  p.  650  (June  1930). 

Stryker,  N.  R.,  Scanning  Losses  in  Reproduction,  VoL  XV,  6,  p.  610  (November  1930). 

Cook,  E.  D.,  The  Aperture  Alignment  Effect,  VoL  XXI,  5,  p.  390  (November  1933). 

Foster,  DM  Effect  of  Orientation  of  the  Scanning  Image  on  the  Quality  of  Sound  Reproduced  from 
Variable  Width  Records,  VoL  XXXIII,  5,  p.  502  (November  1939). 

Batsel,  M.  C.,  and  C.  H.  Cartwright,  Effect  of  Uneven  Slit  Dlumination  on  Distortion  in  Several  Types 
of  Variable  Width  Records,  VoL  XXIX,  5,  p.  476  (November  1937). 

Carlson,  F.  E.t  Properties  of  Lamps  and  Optical  Systems  for  Sound  Reproduction,  VoL  XXXIII, 
1,  p.  80  (July  1939). 

Photographic  Tone  Reprodttctwrn 

Hardy,  A.  C.,  The  Rendering  of  Tone  Values  in  the  Photographic  Recording  of  Sound,  Tram.  Soc. 
Mot.  Pict.  Eng.,  VoL  XI,  31,  p.  475  (September  1927). 

Jones.  L.  A.,  On  the  Theory  of  Tone  Reproduction,  with  a  Graphic  Method  for  the  Solution  of  Prob- 
lems, Vol.  XVI,  5,  p.  568  (May  1931). 

Mees,  C.  E.  K.,  Some  Photographic  Aspects  of  Sound  Recording,  VoL  XXIV,  4,  p.  285  (April  1935). 

Measurement  of  Distortion 
Baker,  J.  O.T  and  D.  H.  Robinson,  Modulated  High  Frequency  Recording  as  a  Means  of  Determining 

Conditions  for  Optimal  Processing,  VoL  XXX,  1,  p.  3  (January  1938). 
Frayne,  J.  G.,  R.  R.  Scoville,  Analysis  and  Measurement  of  Distortion  in  Variable  Density  Recording, 

VoL  XXXII,  6,  p.  648  (June  1939). 

Dimensional  and  Other  Standards 

Motion  Picture  Standards  (Z22)  are  obtainable  from  the  American  Standards  Association,  70  E.  45  St., 
New  York,  N.  Y.,  or  from  the  Society  of  Motion  Picture  Engineers,  342  Madisoa  Aw.,  New  York, 
N.Y. 

PIEZOELECTRIC  CRYSTALS 

By  W.  P.  Mason 

31.  DEFINITION  OF  EFFECTS 

A  piezoelectric  crystal  is  a  crystal  which  suffers  a  change  in  dimension  or  form  propor- 
tional to  an  applied  electrical  potential,  for  small  applied  potentials,  and  conversely  gen- 
erates a  surface  charge  when  subject  to  stresses.  These  properties  of  piezoelectric  crystals 
allow  a  coupling  to  be  made  between  an  electrical  circuit  driving  a  mechanical  circuit  or 


13-56  ELECTROMECHANICAL-ACOUSTIC  DEVICES 

with  the  mechanical  properties  of  the  crystals  themselves  used  to  create  an  electric  voltage 
as  a  result  of  mechanical  motion.  When  a  crystal  is  used  to  couple  an  electrical  to  a 
mechanical  system  it  is  said  to  be  an  electromechanical  transducer.  An  example  of  such  a 
device  is  a  crystal  used  as  the  pickup  unit  in  a  phonograph,  in  which  function  it  transforms 
the  'mechanical  vibrations  of  the  record  into  electrical  vibrations  which  are  amplified  and 
produce  sound  vibrations  through  the  loudspeaker.  When  the  mechanical  resonances  of 
the  crystal  itself  are  used,  the  crystal  is  said  to  be  a  piezoelectric  resonator  or  a  piezoelectric 
oscillator. 

The  piezoelectric  effect  was  discovered  in  1880  by  the  brothers  Jacques  and  Pierre  Curie. 
They  discovered  first  the  "direct"  effect,  which  is  the  production  of  charge  on  a  crystal 
surface  due  to  the  effect  of  a  mechanical  force  applied  to  the  crystal  surface.  They  also 
measured  the  "inverse"  effect,  which  is  the  change  in  shape  of  the  crystal  due  to  an  applied 
potential.  Voigt  (Lehrbuch  der  Kristallphysik,  B.  Teubner,  1910)  later  showed  that  all 
the  linear  properties  of  a  crystal  under  applied  stresses,  potentials,  and  temperatures  re- 
sulted in  strains,  electrical  displacements,  and  increases  in  heat  energy  according  to  the 

equations 

63 

Si  -  S  SijTf  -f  S  dthEk  +  on  BB     (i,  j  =  1,  •  •  -,  6) 

.7=1  Jfc-1 

—  =  23  difTi  +  S  r-  Etc  -f  PI  50     (fc,  I  =  1,  •  •  •,  3)  (1) 

47r       J~i  A=l47r 

6  3 

8Q  =  £  9*/2V  +  2  SpkE*  +  PCP  60 


where  Si(i  —  1  to  6)  are  the  six  strains  that  can  exist  in  a  solid  body,  T3-(j  =  1  to  6)  are 
the  six  stresses  in  the  body,  JSk(k  =  1  to  3)  the  three  potential  gradients  (ratio  of  total 
potential  divided  by  distance  over  which  they  are  applied)  that  exist  along  the  three  axes, 
9  is  the  absolute  temperature  in  degrees  Kelvin  and  59  the  increase  in  temperature, 
Di(l  =  1  to  3)  is  the  electric  displacements  along  the  three  axes,  and  8Q  is  the  increment 
in  heat  energy  due  to  applied  stresses,  fields,  and  temperature  increments.  (These  symbols 
have  now  been  standardized  by  the  Institute  of  Radio  Engineers.) 

The  first  equation  says  that  any  one  of  the  strains,  for  example  Si,  is  in  general  propor- 
tional to  the  six  stresses,  the  three  electric  fields  along  the  three  axes,  and  the  temperature 
increment  56.  The  constants  $»•/  which  relate  the  strains  to  the  applied  stresses  are  the 
elastic  moduli  of  compliance.  Since  it  can  be  shown  that  SH  —  Sji  there  are  21  such  con- 
stants for  the  most  general  crystal,  a  triclinic  crystal.  If  any  elements  of  symmetry  exist 
in  the  crystal  the  number  of  independent  constants  is  reduced.  For  example,  ammonium 
dihydrogen  phosphate  (ADP)  has  six  independent  elastic  compliances,  quartz  has  seven, 
and  Rochelle  salt  (sodium  potassium  tartrate)  has  nine.  The  diu  constants  are  the  piezo- 
electric constants  which  relate  the  strains  to  the  applied  fields.  For  the  most  general 
crystal  there  are  18  independent  constants,  but  for  more  symmetrical  crystals  the  number 
is  reduced.  ADP  has  two  independent  constants,  quartz  two,  and  Rochelle  salt  three. 
The  ai  constants  are  the  six  temperature  expansion  coefficients  which  relate  the  six  strains 
to  the  applied  temperature  increase  56. 

The  second  equation  states  that  the  electric  displacement  is  proportional  to  the  applied 
stresses  (the  constants  of  proportionality  being  again  the  piezoelectric  constants),  to  the 
applied  fields  (the  constants  of  proportionality  being  the  dielectric  constants  e&z)  ,  and  to 
the  increase  in  temperature  56  (the  constant  of  proportionality  being  the  pyroelectric 
constants  pi)  .  (The  equation  as  written  is  valid  for  the  cgs  system  of  units.  For  the  mks 
system  the  4-rr  is  removed  from  Di  and  ejt.)  Since  l/(47r)  times  the  normal  component  of 
the  electric  displacement  at  the  surface  of  the  crystal  is  equal  to  the  surface  charge  <r,  this 
equation  shows  the  origin  of  the  direct  piezoelectric  effect  (while  eq.  [1]  expresses  the  in- 
verse effect). 

The  third  equation  states  that  the  increment  of  heat  8Q  is  proportional  to  the  stresses, 
the  fields,  and  the  applied  temperature  increment.  The  first  effect  is  called  the  stress- 
caloric  effect,  and  the  constant  of  proportionality  is  the  absolute  temperature  6  times  the 
temperature  expansion  coefficients  a,.  The  second  effect  is  called  the  electro  caloric  effect, 
and  the  constant  of  proportionality  is  the  absolute  temperature  6  times  the  pyroelectric 
constant  pk.  This  last  is  the  ratio  of  the  electric  displacement  to  the  applied  temperature 
56  measured  at  constant  stress  and  constant  field.  The  last  term  in  the  third  equation 
expresses  the  increase  in  heat  energy  due  to  a  temperature  increase  59,  and  the  constant 
of  proportionality  is  the  density  p  times  the  specific  heat  at  constant  stress  Cp.  In  all 
these  equations  the  constants  of  proportionality  for  one  variable  are  measured  with  the 


DEFINITION  OF  EFFECTS  13-57 

other  two  variables  in  the  equation  held  constant.  Thus  s,-;,  for  example,  could  be  written 
with  two  superscripts  «f/s.e,  indicating  that  in  determining  the  constants  the  fields  E 
and  the  temperature  6  are  held  constant.  They  are  therefore  the  constant  field,  isothermal 
elastic  compliances.  Similar  superscripts  can  be  written  for  the  other  terms. 

For  most  piezoelectric  applications  the  vibrations  are  so  rapid  that  there  is  no  time  for  an 
interchange  of  ^  heat,  and  adiabatic  conditions  prevail.  This  can  be  taken  account  of  in 
eq.  (1)  by  setting  5Q  =  0.  If  we  do  this  and  eliminate  £6  from  the  remaining  equations 
we  have  two  equations  given  by 

6  3 

st  =  2  *a*Ti  +  2  **#* 
j-i          *=i 

n  6  3         -  (2) 


All  these  constants  are  adiabatic,  and  they  are  related  to  the  isothermal  constants  of  eq.  (1) 
by  the  relations 


where  the  constants  on  the  left  side  are  understood  to  be  adiabatic.  Equations  (2)  repre- 
sent all  the  linear  adiabatic  relations  existing  for  a  piezoelectric  crystal,  while  eqs.  (1)  give 
all  the  isothermal  linear  relations  existing  for  a  piezoelectric  crystal.  Two  second-order 
effects,  the  piezo-optical,  and  the  electro-optical,  are  also  of  some  interest,  but  they  will 
not  be  discussed  here.  They  result  from  a  change  in  the  dielectric  constant  as  a  function 
of  applied  stresses  and  applied  fields  respectively. 

For  ferroelectric  crystals  such  as  Rochelle  salt  the  constants  in  the  equations  of  the 
form  (2)  go  through  very  wide  variations  over  a  temperature  range.  It  has  been  found  * 
that,  if  the  electric  displacement  is  used  as  the  dependent  variable  instead  of  the  field, 
the  resulting  constants  are  nearly  independent  of  temperature.  These  relations  can  be 
obtained  from  eqs.  (2)  by  solving  in  terms  of  DI  and  TJ;  they  are 


6  3 

£*  =   -  2  &>Tj  -f  £  $klTDl  (4) 

.7=1  1=1 

where 

r  —  l)k+i\k,i 
fit?  =  -  -  '—  -  ;        gu  =  fadu  +  ferf23  +  &k*d*i 

-f  rf*4|j4  +  diagfr  -f 


and  A  is  the  determinant 

en     €12 
A  = 

and  Afc'z  is  the  determinant  obtained  by  suppressing  the  &th  row  and  Zth  column. 

A  ferroelectric  crystal  is  one  which  shows  a  spontaneous  polarization  over  a  given 
temperature  range.  This  is  due  to  the  movable  electric  dipoles  exerting  a  mutual  reaction 
and  lining  up  for  one  direction  of  the  crystal.  The  effect  is  similar  to  the  ferromagnetic 
effect  in  magnetic  substances  and  is  accompanied  by  similar  effects.  The  polarization  vs. 
potential  curves  show  hysteresis  effects  and  very  high  dielectric  constants.  Large  piezo- 
electric effects  exist  in  the  ferroelectric  range,  and  Rochelle  salt,  for  example,  has  a  du 
piezoelectric  constant  which  may  be  1000  times  as  large  as  that  for  a  quartz  crystal.  The 
limiting  temperatures  for  the  ferroelectric  regions  are  called  the  Curie  temperatures. 
For  Rochelle  salt  these  are  —18  and  +24  deg  cent.  Other  ferroelectric  crystals  are  also 
known,  notably  potassium  dihydrogen  phosphate  and  potassium  dihydrogen  arsenate  (see 
Busch,  "Neue  Seignette  Elektrica,"  Helv.  Phys.  Ada,  II  No.  3  [1938]),  but  their  Curie 
temperatures  are  very  low,  namely,  — 151  and  — 182  deg  cent.  RocheUe  salt  was  the 
first  crystal  discovered  that  had  its  ferroelectric  region  in  the  room-temperature  range. 
This  fact  accounts  for  its  wide  use  in  acoustic  devices  in  spite  of  its  poor  mechanical  and 

*  See  W.  P.  Mason,  A  Dynamic  Measurement  of  the  Elastic,  Electric,  and  Piesoelectric  Constants 
of  RocheUe  Salt,  Phys.  Rev,,  Vol.  55,  775  (1939),  and  H.  Mueller,  Properties  of  Rochelk  Salt,  Pk&s. 
Rev.,  57,  829  (1940). 


13-58  ELECTROMECHANICAL-ACOUSTIC  DEVICES 

chemical  properties.  Another  ferroelectric  crystal,  barium  titanate,  has  recently  been 
discovered  which  has  a  large  electrostrictive  effect.  This  crystal  in  ceramic  form  may 
be  an  important  electromechanical  transducing  element  (Mason,  "Electrostrictive  Effect 
in  Barium  Titanate  Ceramics,"  Phys.  Rev.,  Vol.  74,  No.  9,  pp.  1134-1148,  Nov.  1,  1948). 

32.  APPLICATION  OF  PIEZOELECTRIC  CRYSTALS 

The  piezoelectric  effect  remained  a  scientific  curiosity  from  the  time  of  its  discovery 
until  the  time  of  World  War  I,  1914-1918.  During  that  time  Professor  Langevin  in  Paris 
devised  an  underwater  sound-locating  device  using  quartz  crystals  to  convert  alternating 
electrical  energy  into  sound  vibrations  in  water.  The  sound  beam  was  sent  out  into  the 
water  and  was  reflected  back  from  an  object  or  the  bottom  of  the  ocean.  This  reflection 
impinged  on  the  crystal  transducer  and  generated  an  electrical  voltage  which  could  be 
detected  by  vacuum-tube  devices.  This  use  was  a  forerunner  of  the  fathometers  and 
•underwater  sound-locating  devices  that  have  been  widely  used  by  the  Navy.  Although 
quartz  was  originally  used  for  this  purpose  it  has  been  displaced  by  Rochelle  salt  and 
particularly  a  new  crystal  developed  during  World  War  II,  the  ammonium  dihydrogen 
phosphate  or  ADP  crystal.  This  crystal  has  so  many  mechanical,  chemical,  power- 
handling  capacity,  and  temperature  advantages  over  Rochelle  salt  that  it  appears  likely 
to  replace  all  other  transducing  elements  for  underwater  sound  applications. 

In  1922  it  was  shown  by  Professor  Cady  of  Wesleyan  University  that  very  stable 
oscillators  could  be  obtained  by  using  quartz  crystals  as  the  frequency-controlling  element. 
These  have  been  applied  to  controlling  the  frequency  of  broadcasting  stations  and  radio 
transmitters  in  general.  Quartz  crystals  using  some  one  of  the  low-temperature-coefficient 
crystals  described  in  article  33  produce  the  most  stable  oscillators  and  the  best  time- 
keeping systems  that  can  be  obtained.  The  use  of  crystals  to  stabilize  oscillators  was  so 
prevalent  during  World  War  II  that  over  30,000,000  crystals  were  produced  in  a  single 
year  for  this  purpose. 

Another  application  of  piezoelectric  crystals  is  in  producing  very  selective  filters.  On 
account  of  the  very  high  Q  existing  in  crystals  they  can  practically  eliminate  the  effect 
of  dissipation  in  filter  structures.  Such  filters  have  been  widely  applied  in  the  long-dis- 
tance telephone  lines  and  in  single-sideband  transatlantic  radio  telephone  systems. 
Narrow-band  crystal  filters  have  been  used  in  picking  off  single  frequencies  and  narrow 
bands  of  frequencies  for  control  and  analyzing  purposes.  For  this  application  quartz 
crystals  have  been  mostly  used.  However,  it  appears  that  the  requirements  are  lenient 
enough  to  allow  some  of  the  synthetic  crystals  to  be  employed. 

Besides  producing  and  detecting  sound  in  liquids,  crystals  have  been  used  to  produce 
and  detect  vibrations  in  gases  and  solids.  On  account  of  their  high  mechanical  and 
electrical  impedances  crystals  are  at  somewhat  of  a  disadvantage  in  coupling  to  low- 
mechanical-impedance  air  waves.  By  using  bimorph  types  of  units  which  employ  bending 
or  flexnral  vibrations  the  mechanical  impedances  of  crystals  can  be  lowered.  For  sound 
pickup  devices  the  high  electrical  impedance  is  not  a  disadvantage,  for  they  can  be  worked 
directly  into  the  grid  of  a  vacuum  tube  which  inherently  is  a  high  impedance.  Hence 
large  numbers  of  crystals  have  found  uses  in  microphones.  For  this  purpose  Rochelle  salt 
is  common,  but  the  constants  of  ADP  are  favorable  enough  so  that  they  may  displace  it. 

Crystals  have  also  been  used  in  receivers,  relays,  oscillographs,  and  other  devices  for 
which  displacements  are  required  for  a  given  applied  voltage.  For  this  purpose,  crystals 
having  large  d  piezoelectric  constants  are  required,  and  Rochelle  salt  is  universally  used. 
On  account  of  the  large  variation  of  d  with  temperature  such  devices  are  not  very  stable 
and  reproducible  and  hence  are  unsuitable  for  high-quality  equipment. 

Crystals  have  also  been  employed  in  producing  very  high-frequency  vibrations  in  gases, 
liquids,  and  solids.  For  this  purpose  quartz  is  the  almost  universal  choice  since  it  can  be 
ground  very  thin  and  can  be  used  to  produce  high  frequencies.  X-cut  quartz  is  utilized 
to  set  up  longitudinal  vibrations  and  Y-cut  quartz  to  produce  shear  vibrations.  Such 
high-frequency  sound  waves  are  applicable  for  testing  steel  castings  and  other  solid 
materials  for  flaws  (Firestone,  The  Supersonic  Reflectoscope,  .7".  A.S.A.,  Vol.  17,  No.  3 
[January  1946]).  They  have  also  been  utilized  to  study  the  properties  of  liquids,  gases, 
and  solids  and  the  way  in  which  they  vary  with  frequency. 

33.  PROPERTIES  OF  QUARTZ 

There  are  at  least  500  crystalline  substances  that  have  been  tested  and  found  to  be 
piezoelectric,  and  it  is  to  be  presumed  that  among  the  many  thousands  of  compounds  that 
will  form  into  crystals  in  the  20  out  of  the  32  crystallographic  classes  that  may  be  piezo- 


PROPERTIES  OF  QUARTZ 


13-59 


electric  most  of  them  will  show  some  piezoelectric  activity.  However,  only  three  crystals 
nave  received  wide  application  in  practical  devices:  quartz.  Rochelle  salt,  and  ammonium 
dihydrpgen  phosphate  (ADP).  A  fourth  crystal,  tourmaline,  has  received  a  limited  appli- 
cation in  sound-measuring  devices  because  it  is  sensitive  to  hydrostatic  pressures.  It  is  to  be 
expected,  however,  that,  with  several  large  laboratories  actively  engaged  in  investigating 
new  piezoelectric  materials,  many  more  crystals  will  eventually  find  practical  appHcation. 
It  is  the  purpose  of  the  following  sections  to  discuss  the  properties  and  useful  cuts  of 
quartz,  Rochelle  salt,  and  ADP. 

PHYSICAL  PROPERTIES  OF  QUARTZ.  Quartz  is  described  by  the  chemist  as 
silicon  dioxide,  Si02,  and  it  crystallizes  in  the  trigonal  trapezohedral  class.  The  Z  or  optic 
axis  is  an  axis  of  threefold  symmetry;  i.e.,  if  one  measures  any  property  of  the  crystal  at  a 
definite  position  in  the  crystal,  this  property  will  be  repeated  at  angles  of  ±120°  rotation 
about  the^Z  axis.  The  melting  point  of  quartz  is  1750  deg  cent,  the  density  2.65,  and  the 
hardness  is  7  on  Mohs'  scale.  Under  atmospheric  pressure,  a  or  low-temperature  quartz 
transforms  into  ft  or  high-temperature  quartz  at  573  deg  cent.  Under  stress  this  trans- 
formation temperature  is  lowered.  Alpha  quartz  is  insoluble  in  ordinary  acids  but  is 
decomposed  in  hydrofluoric  acid  and  in  hot  alkalies.  Quartz  is  soluble  to  some  extent  in 
water  at  high  pressures  and  temperatures.  In  an  enclosed  system,  crystalline  quartz  will 
dissolve  in  water  to  the  extent  of 
3  grams  per  liter  at  350  deg  cent. 
Powdered  fused  quartz,  which  has 
a  larger  surface-to-volume  ratio, 
will  dissolve  to  a  considerably 
larger  extent. 

Quartz  is  found  principally  in 
Brazil  in  several  different  types  of 
deposits  (see  Stoiber,  Tolman,  and 
Butler,  Geology  of  Quartz  Crystal 
Deposits,  Am.  Mineralogist,  Vol. 
30,  245-268  [1945]).  The  prepon- 
derance of  the  crystals  found  is  in 
the  lower-weight  class  as  shown  by 
the  table.  Most  of  the  clear  quartz 
has  recognizable  natural  faces,  but 
some,  particularly  river  quartz, 
has  no  natural  faces. 

Quartz  occurs  in  optical  right-hand  and  left-hand  forms;  i.e.,  the  crystal  will  rotate  the 
plane  of  polarization  of  polarized  light  passing  along  the  Z  or  optic  axis  counterclockwise 
(left  handed)  or  clockwise  (right  handed)  from  the  point  of  view  of  the  observer  facing  the 
source  of  light.  Most  crystals  have  sections  with  both  handedness.  In  general,  the  middle 
section  is  likely  to  be  all  of  one  hand  while  the  outside  sections  may  have  parts  of  each 
handedness.  A  conoscope  may  be  used  to  locate  the  optic  axis  and  will  also  show  the 
handedness  and  position  of  any  optical  twinning.  The  principle  of  the  conoscope  is  shown 
by  Fig.  1.  Light  from  the  source  is  sent  through  a  polarizer  aad  through  the  converging 
lens  LI.  This  lens  sends  converging  or  conical  beams  through  the  crystal  which  are 
gathered  by  the  second  lens,  focused,  and  sent  through  the  analyzer.  In  practice  the 
lenses  and  crystal  are  immersed  in  a  liquid  having  the  same  index  of  refraction  as  the  crystal 

along  its  optic  axis.  Such  liquids 
may  be  mixtures  of  Decaliii  and 
Dowtherm  or  dimethyl  phthalaie 
and  a  monochlor  naphthalene. 
The  crystal  breaks  up  all  rays  not 
parallel  to  the  optic  axes  into  two 
components  which  travel  with  dif- 
ferent velocities.  Hence  the  an&- 
FIG.  1.  Principle  of  Conoscope  lyser  »  not  able  to  extinguish  the 

light  that  has  traversed  the  crystal 

except  at  angles  for  which  the  two  rays  are  in  opposite  phase.  Hence  one  sees  a  series  of 
rings  in  the  conoscope  when  the  direction  of  the  Z  axis  is  along  the  line  between  the 
source  and  the  eye.  Owing  to  the  rotation  of  the  plane  of  polarization  in  the  crystal  one 
finds  that  the  rings  either  expand  or  contract  for  a  right-  or  left-handed  crystal  respectively 
for  a  clockwise  rotation  of  the  analyzer.  This  gives  a  method  of  determining  the  handed- 
ness of  the  crystal.  Optical  twinning  also  shows  up  in  a  viewing  system  of  this  type,  for 
it  deforms  the  ring  pattern.  If  plane  rays  rather  than  conical  rays  are  used,  and  a  source 
of  white  light,  color  effects  also  show  up  the  position  of  the  optical  twinning. 


Crystal  Weight  Groups 
weight  in  grams 

Percentage  of  the  Total  Number 
of  Crystals  Which  Were  In  Each 
Weight  Group 

200-       300 
3Q&-       50Q 
500-      700 

55.5 
29.5 
I©.  4 

700-  !,000 
1,000-  2,000 
2,000-  3,000 

2.1 

1.  8 

0.5 

3rOOO-  4,006 
4,000-  5,000 
5,000-  7,000 

0.2 
<0.1 
<0.1 

7,000-10,000 

<Q,f 

13-60  ELECTROMECHANICAL-ACOUSTIC  DEVICES 


NORMAL  ALPHA  QUARTZ 


The  Dauphine  or  electrical  type  of  twinning  also  exists  in  quartz.  It  results  from  a  180° 
change  in  the  direction  of  the  crystal  atomic  arrangements.  As  shown  by  Fig.  2,  the  silicon 
atoms  normal  to  the  Z  axis  are  arranged  in  near  hexagons  all  pointing  in  one  direction. 
If  the  temperature  is  raised  above  573  deg  cent  a  change  in  the  arrangement  to  the  hexag- 
onal pattern  shown  in  the  middle  occurs.  The  result  is  high-temperature  or  /3  quartz. 
As  the  temperature  is  decreased  below  573  deg,  the  crystal  may  return  to  the  form  at 
the  bottom,  or  part  of  it  may  return  to  this  form  and  part  to  the  form  in  which  the  near 
hexagons  point  in  the  opposite  direction.  If  both  forms  exist  the  crystal  is  said  to  have 

electrical  twinning.  The  best  method  of  detecting  elec- 
trical twinning  is  by  etching  the  surface  with  hydrofluoric 
acid,  which  eats  away  the  crystal  at  rates  depending  on 
the  orientation  of  the  crystal  surface.  Since  the  two 
twinned  areas  will  develop  etch  pits  that  point  in  op- 
posite directions,  grazing  light  will  cause  one  part  to  re- 
flect brightly  while  the  other  reflects  diffusely,  and  hence 
one  can  see  the  parts  that  have  different  regions  of  elec- 
trical twinning.  Since  the  piezoelectric  effect  is  opposite 
for  the  two  twinned  areas,  it  is  necessary  that  there  be 
only  one  region  in  a  useful  crystal.  Electrical  twinning 
usually  occurs  in  an  untwinned  plate  if  it  is  taken  above 
the  inversion  point.  It  may  also  occur  at  lower  temper- 
atures if  stress  is  applied.  Such  twinning  has  been  ob- 
served when  a  hot  soldering  iron  is  pressed  against  a 
crystal,  or  it  may  even  occur  when  the  crystal  is  sawed. 
Wooster  (Nature,  Vol.  157,  No.  3987  [March  30,  1946]) 
has  found  that  the  electrical  twinning  can  be  removed  by 
exerting  a  twist  around  the  Z  or  Z'  axis  and  heating  the 
crystal  nearly  to  the  inversion  point. 

Other  defects  in  quartz  crystals  are  (1)  "bubbles": 
bubble-like  cavities  which  may  be  fine  or  large;  (2)  veils, 
heavy  or  fine,  which  are  more  or  less  continuous  sheets 
of  small  bubble-like  cavities;  (3)  clouds  or  haze:  aggre- 
gates of  fine  bubble-like  cavities;  (4)  ghosts  or  phantoms: 
outlines  of  earlier  growths  within  the  crystal,  usually 
marking  what  were  once  edges  of  adjoining  faces  which 


xxxx 


BETA  OR  HIGH -TEMPERATURE 
QUARTZ 


TWINNED  ALPHA  QUARTZ 

PIG.    2.      Arrangement    of    Silicon 
Atoms  in  Quartz  Normal  to  Z  Axis 


become  visible  when  a  beam  of  light  is  reflected  from  the  minute  fractures  or  parting 
planes  that  outline  them;  (5)  fractures.  All  these  defects  can  be  observed  by  shining  a 
strong  light  through  the  crystal  at  right  angles  to  the  direction  of  observation.  The 
crystal  is  usually  immersed  in  an  inspection  tank  which  is  filled  with  a  liquid  having  the 
same  index  of  refraction  as  the  crystal.  Opinions  differ  on  how  many  inclusions  or 
bubbles  of  a  small  size  can  be  tolerated  in  the  finished  crystal. 

All  the  inspection  and  orienting  instruments  as  well  as  the  methods  of  sawing  and  pre- 
paring the  crystals  are  completely  described  in  R.  A.  Heising,  Quartz  Crystals  for  Electrical 
Circuits,  Van  Nostrand,  1946,  and  in  the  May-  June  1945  issue  of  the  American  Mineralogist. 

USEFUL  CRYSTAL  ORIENTATIONS.  The  modes  of  motion  and  the  properties  of 
these  modes  depend  markedly  on  how  crystal  plates  are  oriented  with  respect  to  the 
natural  crystal  faces.  Figure  3  shows  a  natural  quartz  crystal,  the  three  crystallographic 
axes,  and  some  of  the  more  important  special  cuts  that  have  found  use  in  the  radio  and 
telephone  art.  The  Z  or  optic  axis  of  the  crystal  is  along  the  long  direction  of  the  crystal; 
the  X  axis  lies  through  one  of  the  apexes  of  the  hexagon;  and  the  T  axis  is  normal  to  the 
other  two  in  a  right-handed  system.  The  piezoelectric,  elastic,  and  dielectric  equations 
of  quartz  take  the  form 

Si    =    811*3*1 


duEx 


-f 


84  = 

Ss  = 


(5) 


47T 


<?z   =   —    — 


PROPERTIES  OF  QUAJRTZ 


13-61 


ZERO 

TEMPERATURE-COEFFICIENT 
OSCILLATORS  AND  FILTERS 

LOW-FREQ1 


HIGH-FREQ: 
AT  (+35°'15') 
BT  (-49°) 


DT   (-52°! 
ET  (+&6°) 
*        FT  (-57°) 


\ 


S&--TV 


ZERO  COUPLING  (Sa4=0) 

-13*  FILTERS 
-3-5-7  HAF*4ONltCS 


0°  OSCILLATORS      \ 
FUNDAMENTAL  AND      * 

SECOND  HARMONIC 


MT  LONCTTUOtMAl. 

CRYSTAL 
NT  FLEXURE   CRYSTAL 


LOW  Te-*PERATUR£- COEFFICIENT 
OSCILLATORS         *      *  +5°  FILTERS 

AND  FILTERS          DOUGHNUT  . 


ZERO  TEMPERATURE-COCFFICIENT 

FIG.  3.     Principal  Cuts  of  Quartz 

where  £1,  &,  S*  are  the  three  elongation  strains  along  the  X,  F,  and  Z  axes  respectively; 
S4,  S6,  and  S6j  the  three  shearing  strains;  jfi,  T2,  T3,  the  three  tensional  stresses;  T*,  T^ 
and  TS,  the  three  shearing  stresses;  Ex,  Eyj  ESl  the  three  fields;  Dx,  Dyj  Z>2,  the  three  elec- 
trical displacements  which  at  the  outer  surfaces  are  equal  to  the  surface  charge  4iro-a., 
47ro-y,  and  47rcrz.  In  cgs  units  the  elastic,  piezoelectric,  and  dielectric  constants  have  the 
values  (see  Mason,  "Quartz  Crystal  Applications,"  B.S.TJT.,  VoL  7TX7T,  No.  2  {July 
1943]) : 

«ii*  =  127.9  X  10~14  X  cmVdyne 

SnE  =  -15.35 


'  =    -11.0 

1  =  -44.6 
;  -  95.6 
'  =  197.8 
=  2(snE  -  sis*)  =  286.5  X  10  ~14 

«  —6.76  X  10~8  statcoulomb/dyne 
=  2.56  X  10~8 


(6) 


4.5S  : 
4.70 


statcoulomb 
statvolt 


for  the  nxks  system  the  elastic  compliances  are  multiplied  by  10,  the  piezoelectric  constants 
divided  by  30,000,  and  the  dielectric  constants  multiplied  by  the  factor  S.S5  X  10"12. 


13-62  ELECTROMECHANICAL-ACOUSTIC  DEVICES 

X-CTJT  CRYSTALS.  These  equations  are  useful  in  predicting  the  type  of  motion 
that  will  be  generated  in  a  given  type  of  cut  and  the  magnitude  of  the  electromechanical 
coupling.  For  example,  the  first  equation  of  eq.  (5)  shows  that  a  strain  Si,  which  is  an 
elongation  along  the  X  axis,  will  be  generated  by  a  field  applied  along  the  X  axis.  The 
applied  field  will  then  generate  a  thickness  longitudinal  mode  since  the  motion  is  in  the 
same  direction  as  the  applied  field.  If  the  thickness  is  made  small  this  type  of  crystal  can 
produce  a  very  high  frequency,  and  it  was  originally  used  to  control  oscillators.  Because 
of  their  poor  temperature  coefficient,  such  crystals  have  largely  been  replaced  in  the 
control  of  oscillators  by  AT  and  BT  thickness  shear  mode  crystals  which  have  much 
better  properties.  X-cut  crystals,  however,  will  produce  ultrasonic  vibrations  in  solids, 
liquids,  and  gases,  and  such  waves  have  been  used  in  studying  the  properties  of  these 
materials  and  also  in  flaw  detectors  (see  Firestone,  /.  A.S.A.  [January  1946]}  which  deter- 
mine whether  any  cracks  or  irregularities  occur  in  metal  castings.  For  this  purpose  it  is 
desirable  to  transform  as  much  input  electrical  energy  as  possible  into  mechanical  energy. 
A  measure  of  the  efficiency  of  this  conversion  for  statically  or  slowly  varying  applied  fields 
is  the  electromechanical  coupling  factor  k,  which  is  defined  by  the  equation 

fc  =  <W^T-  =  °'095  (7) 

*     Ei 

where  CHE  is  the  effective  elastic  constant  for  a  thickness  mode.    This  is  equal  to 

CUE  =  8.60  X  10U     dynes  per  cm2  (8) 

Inserting  the  values  given  in  eq.  (7),  we  find  that  the  coupling  is  about  9.5  per  cent.  This 
means  that,  for  a  static  field,  the  square  of  k  or  about  1  per  cent  of  the  input  energy  is 
stored  in  mechanical  form.  For  alternating  fields  near  the  resonance  of  the  crystal  a 
considerably  larger  part,  in  fact,  nearly  all,  can  be  converted  into  mechanical  energy  if 
the  shunt  capacity  is  tuned  by  a  coil,  but,  nevertheless,  the  coupling  is  a  measure  of  the 
width  of  the  frequency  range  for  which  this  conversion  can  be  done  efficiently.  If  /g  is 
the  highest  frequency  and  /A  the  lowest  frequency  for  which  the  loss  is  not  more  than  50 
per  cent  it  can  be  shown  that 

/~          /i  _i_  z> 

(9) 

Some  synthetic  crystals  such  as  lithium  sulfate  and  L-cut  Rochelle  salt  have  coupling 
factors  of  0.35  to  0.4  and  are  to  be  preferred  when  it  is  desired  to  radiate  a  wide  band  of 
frequencies,  but  for  high  frequencies  X-cut  quartz  is  commonly  used  on  account  of  its 
excellent  mechanical  properties. 

The  second  equation  of  (5)  shows  that  a  strain  $2,  which  is  an  elongation  along  the  Y 
axis,  is  excited  when  a  field  is  applied  along  the  X  axis.  Since  the  long  direction  of  the 
crystal  is  taken  along  this  direction  this  mode  of  motion  is  called  a  length  longitudinal 
mode.  It  has  been  used  to  some  extent  to  drive  low-frequency  oscillations  in  gases,  liquids, 
and  solids.  Two  modifications  of  this  cut  have  received  considerable  use  in  the  construc- 
tion of  quartz  crystal  filters.  These  cuts  are  the  — 18°  -ST-cut  crystal  and  the  +5°  X-cut 
crystal  shown  by  Fig.  3.  The  -—18°  cut  is  used  because  it  produces  a  very  pure  frequency 
spectrum  giving  only  a  single  resonance  over  a  frequency  range  of  3  to  1  (see  W.  P.  Mason, 
Electrical  Wave  Filters  Employing  Quartz  Crystals  as  Elements,  B.S.T.J.,  Vol.  XIII 

[July  1934]}.  The  +5°  .XT-cut  crystal  is 
used  because  it  is  the  best  orientation  of 
the  X  cuts  for  giving  a  low  temperature 
coefficient  of  frequency.  By  putting  a 
divided  plating  on  the  crystal  as  shown  by 
Fig.  4  this  crystal  can  be  driven  in  a  flexure 
mode  at  much  lower  frequencies  than  can  be 
realized  with  a  longitudinal  mode.  It  has 
FIG.  4.  Plating  Arrangement  for  Driving  a  keen  usec*  *or  picking  off  single-frequency 
Longitudinal  Crystal  in  Flexure  pilot  channels  for  controlling  the  gain  of  a 

carrier  system. 

The  temperature  coefficient  of  the  +5°  X-cut  used  for  both  longitudinal  and  flexure 
modes  can  be  improved  by  rotating  the  thickness  around  the  length  of  the  crystal.  This 
results  in  the  M T  and  NT  crystals  shown  by  Fig.  3.  These  have  temperature  coefficients 
under  one  part  in  a  million  per  degree  centigrade  but  a  smaller  coupling  than  the  equivalent 
-1-5°  X-cut  crystals.  (See  Mason  and  Sykes,  Low  Frequency  Quartz  Crystal  Cuts  Having 
Low  Temperature  Coefficients,  Proc.  LR.E.,  Vol.  32,  No.  4  [April  1944].) 

F-CUT  CRYSTALS.  When  a  field  Ev  is  applied  along  the  Y  axis,  eq.  (5)  shows  that 
two  types  of  strain  are  generated,  S&  and  SQ.  Both  these  strains  are  shearing  strains  which 


PEOPERTIBS  OF  QUARTZ 


13-63 


distort  a  square  in  the  crystal  into  a  rhombus  as  shown  by  Fig.  5.  The  Sg  strain,  shown 
in  Fig.  5,  distorts  the  crystal  in  the  XZ  plane;  the  &  strain  distorts  the  crystal  in  the  XY 
plane.  ^  Since  the  field  is  applied  along  the  thickness, 
which  is  the  Y  direction,  the  first  strain  5*  is  called  a 
face  shear  strain  and  Ss  a  thickness  shear  strain.  The 
frequency  of  a  face  shear  mode  is  controlled  by  the  con- 
tour dimensions  and  hence  will  be  relatively  low.  The 
frequency  of  the  thickness  shear  mode  is  controlled  by 
the  thickness  dimension]  which  can  be  made  very  small 
and  hence  will  result  in  a  high  frequency. 

The  7-cut  crystal  was  first  used  in  the  control  of  high- 
frequency  oscillators  but  on  account  of  its  high  temper- 
ature coefficient  has  largely  been  displaced  by  the  AT 
and  BT  crystals  which  are  modified  F-cut  crystals.  The 
7-cut  crystal  is  still  used  to  generate  shear  vibrations  in  i  ^X 

solids.   For  this  purpose  it  has  a  higher  coupling  than  the     /! ^ 

X  cut,  since  the  coupling  for  the  shear  thickness  mode  is    *• — " 


0.142 


FIG.  5.  Method  for  Obtaining  a 
,  „  rt  Longitudinal  Vibration  from  a  Shear 
(10)  Crystal 


FREQUENCY  CONSTANT  IN  KILOCYCLt 
MILLIMETERS  (Fxd) 

«  s  8  B  K  8 

—^e. 

/ 

--— 

f* 

BC 

>> 

Vk 

/ 

s. 

\ 

/ 

\ 

\ 

I/ 

N 

I^Y-CUT 

/ 

/ 

\ 

AC    XT  S 

H< 

-6O         -60-40-20           0            20           4O          6O          8O 

ROTATION   ABOUT  X  AXIS  iN  DECREES  (9) 
FIG.  6.     Frequency  Constants  for  Rotated  F-cut  Quartz  Crystals 


Rotations  of  the  thickness  direction  around  the  X  axis  have  resulted  in  rotated  Y  cuts 
that  have  very  favorable  properties.     Investigations  made  by  Lack,  Willard  and  Fair, 
,  Koga,    Bechmann,    and 

Straubel  have  shown 
how  the  properties  of  the 
thickness  shear  mode 
varied  with  angle  of  cut. 
As  shown  by  Fig.  3  all 
the  orientations  result- 
ing in  useful  crystals 
have  their  length  along 
the  X  axis  and  their 
thickness  makes  an  angle 
&  with  the  Y  axis.  Fig- 
ure 6  shows  the  fre- 
quency constant  (kilo- 
cycles for  a  crystal  1  Tnm 
thick)  as  a  function  of 
the  angle  of  rotation. 
At  an  angle  of  rotation 
of  -f  31°  and  -59°  the 

frequency  is  minimum  and  maximum  respectively.  At  these  two  angles,  the  mechanical 
coupling  between  the  thickness  shear  mode,  and  the  face  shear  mode  and  overtones,  be- 
comes zero  and  a  crystal  is  obtained  which  is  much  freer  from  extraneous  modes  of  motion 
than  is  the  Y  cut.  Fig- 
ure 7  shows  a  plot  of 
temperature  coefficient 
against  the  orientation 
angle,  and  at  35°  15'  and 
—  49°  crystals  are  ob- 
tained which  have  zero 
temperature  coefficients. 
These  cuts,  known  as  the 
AT  and  BT  crystals  re- 
spectively, have  been 
very  widely  used  to  con- 
trol high-frequency  oscil- 
lators. Frequencies  as 
high  as  10  megacycles 
are  used  for  fundamental 
control,  and  by  means 
of  mechanical  harmonics 
frequencies  as  high  as  197 
megacycles  have  been 


TEMPERATURE  COEFFICIENT  IN  PARTS 
,  PER  MILLION  PER  *C 

§  8  i  o  fc  8  § 

0  EXPERIMENTAL 
CHECK   POINTS 

^i 

/ 

/ 

\ 

\ 

BT 

/ 

/ 

\ 

.AC 

V3! 

BO 

/ 

> 

s, 

/ 

/ 

V 

\ 

S 

L>*«>. 

—  •* 

-80 

-60         -40       -20             0            2O          4O           60          8O 
ROTATION   ABOUT  X  AXIS  JN  DEGREES  (6) 

FIG.  7.    Temperature  Coefficients  for  Rotated  F-cut  Quartz  Crystals 


13-64  ELECTROMECHANICAL-ACOUSTIC  DEVICES 

obtained.  (See  Mason  and  Fair,  A  New  Direct  Crystal  Controlled  Oscillator,  Proc.  I.R.E., 
Vol.  30,  464-472  [October  1942].) 

Since  the  AT  and  BT  are  relatively  near  in  angle  to  the  AC  and  BC  cuts  they  have  a 
good  frequency  spectrum.  Strong  couplings  still  exist  to  flexure  modes  of  motion.  By 
measuring  the  modes  of  motion  as  a  function  of  the  length,  width,  and  thickness,  dimen- 
sional ratios  can  be  obtained  for  which  only  the  main  mode  exists  for  a  large  frequency 
range  on  either  side  of  the  main  frequency.  (See  Sykes,  Modes  of  Motion  in  Quartz 
Crystals,  B.S.T.J.,  Vol.  XXIII,  No.  1  [January  1944J.)  By  maintaining  this  ratio  fixed 
as  the  thickness  is  changed,  a  good  crystal  free  from  resonances  over  a  wide  temperature 
range  is  obtained.  Crystals  produced  by  the  process  of  grinding  to  a  set  of  predetermined 
dimensions  are  called  predimensioned  crystals  and  usually  result  in  a  higher-activity 
crystal  and  one  having  a  smooth  temperature  frequency  curve  over  a  wide  temperature 
range. 

Another  manufacturing  process  called  the  edge  grinding  process  is  sometimes  employed. 
This  consists  in  controlling  the  thickness  dimension  only  and  in  removing  troublesome 
couplings  by  grinding  the  edges  of  the  crystal  until  the  crystal  has  a  high  activity  and  is 
free  from  frequency  hops  over  a  temperature  range.  This  process  may  be  quicker  for 
crystals  that  do  not  have  to  satisfy  stringent  activity  and  temperature  requirements  but 
is  not  likely  to  produce  as  satisfactory  crystals  as  the  predimensioning  process.  Thickness 
vibrating  crystals  may  either  be  ground  or  etched  to  frequency.  On  account  of  an  aging 
which  appears  to  be  due  to  loosely  bound  and  misoriented  layers  of  quartz  on  the  surface 
caused  by  sawing  and  lapping  processes,  it  has  become  customary  to  etch  crystal  surfaces 
to  frequency,  since  this  process  removes  the  loosely  bound  material  and  leaves  a  surface 
that  does  not  age  appreciably.  The  aging  appears  to  be  caused  by  the  attack  of  water 
vapor  on  the  strained  surface  which  results  in  either  loosening  or  removing  the  strained 
material.  The  first  process  causes  a  lowering  of  the  Q  of  the  crystal  (ratio  of  reactance  to 
resistance)  and  a  consequent  lowering  of  the  activity  of  the  oscillator  controlled  by  the 
crystal;  the  second  process  causes  an  increase  in  the  frequency  of  the  crystal.  Aging  can 
be  prevented  by  etching  the  crystal  surface  to  a  depth  of  several  microns  or  by  hermetically 
sealing  the  crystal. 

Two  other  methods  of  adjusting  the  frequency  of  crystals  have  been  employed.  One 
(see  Sykes,  High  Frequency  Plated  Quartz  Crystal  Units  for  Control  of  Communications 
Equipment,  Proc.  I.R.E.,  Vol.  34,  No.  2  [February  1946])  etches  the  crystal  frequency 
above  the  desired  frequency  by  a  predetermined  number  of  kilocycles  and  then  lowers 
the  frequency  by  plating;  by  an  evaporation  process  an  amount  of  metal  necessary  to 
load  the  crystal  down  to  its  desired  frequency  is  added  to  the  crystal.  By  this  method  the 
frequency  can  be  very  accurately  controlled  in  the  final  mounting.  The  other  method 
utilizes  the  recently  discovered  fact  that  exposure  to  X-ray  irradiation  lowers  the  elastic 
constant  of  BT  and  AT  crystals  and  hence  lowers  their  frequency  of  oscillation  (see 
Frondel,  Effect  of  Radiation  on  the  Elasticity  of  Quartz,  Am.  Mineralogist,  Vol.  30  [May 
1945]).  The  effect  is  produced  by  electrons  being  expelled  from  orbits  around  silicon 
atoms  in  the  quartz  and  causing  a  lower  energy  of  binding  between  molecules  and  hence  a 
slightly  lower  elastic  constant.  This  effect  amounts  to  0.1  per  cent  frequency  change  at 
the  most  and  varies  by  considerable  factors  from  crystal  to  crystal,  presumably  owing  to 
the  amount  of  their  impurity  content.  Exposure  to  X-rays  causes  a  darkening  of  the 
crystal,  and  the  amount  of  darkening  appears  to  be  correlated  with  the  amount  of  fre- 
quency change.  On  account  of  the  variability  of  the  effect,  this  process  has  not  had  a 
wide  use. 

Two  other  rotated  F-cut  crystals  that  can  be  given  zero  temperature  coefficients  are 
the  CT  and  DT  face  shear  cuts  (see  Willard  and  Hight,  Proc.  I.R.E.,  Vol.  25,  549-563 
[1937]) .  These  are  nearly  at  right  angles  to  the  AT  and  BT  cuts  and  use  the  same  shearing 
moduli  in  the  face  shear  mode  that  the  AT  and  BT  do  in  the  thickness  shear  mode.  The 
CT  cut  at  +38°  orientation  as  shown  by  Fig.  3  has  a  frequency  constant  of  308  kc-cm  for  a 
square  crystal  and  has  been  used  in  frequency-modulated  oscillators  in  the  frequency  range 
from  300  to  1000  kc.  The  DT  crystal  is  smaller  for  the  same  frequency  and  is  used  in  the 
frequency  range  from  200  to  500  kc.  Both  these  crystals  had  wide  application  in  frequency- 
modulated  oscillators  for  tank  and  artillery  radio  circuits  during  World  War  II. 

The  final  rotated  F-cut  crystal  that  has  been  used  considerably  for  controlling  very 
precise  oscillators  for  time  standards  and  in  the  Loran  navigation  system  is  the  GT  crystal 
(see  Mason,  A  New  Quartz  Crystal  Plate,  Designated  the  GTT  Proc.  I.R.E.,  Vol.  28,  20- 
223  [May  1940]).  This  crystal  is  produced,  as  shown  by  Fig.  3,  by  rotating  the  plane 
by  51°  7.5'  from  Y  and  by  rotating  the  length  45°  from  the  X  axis.  Whereas  most  other 
zero-temperature-coefficient  crystals  have  a  parabolic  variation  of  frequency  with  tem- 
perafenare  about  the  zero  temperature  coefficient  as  shown  by  Fig.  8,  this  parabolic  variation 
is  absent  for  the  GT  and  a  very  constant  frequency  is  produced  over  a  wide  temperature 


PROPERTIES  OF  ROCHELLB  SALT 


13-65 


range.  Hence  a  very  moderate  temperature  control  produces  a  very  constant  frequency. 
A  crystal  mounted  by  means  of  several  wires  soldered  to  its  surface  (see  Greenidge,  Mount- 
ing and  Fabrication  of  Plated  Quartz  Crystal  Units,  B.S.T.J.,  Vol.  23,  234  [July  1944J) 


20 


z 
o 


a 
£ 


50 


/      /LONG  BAR»  LENGTH  \ 

/^Y        ALONG  X  AXIS:       \ 

/      )L>1ST   HARMONIC       \ 

/        /    ^~2ND  HARMONIC         V 


V 


/v 

/A 


t 


\ 


\ 


\\ 


\ 


10  2O  30  40  5O  60  70  »O  9O  1OO 

TEMPERATURE  IN  DECREES  CENTIGRADE 
G.  8.     Temperature  Frequency  Characteristics  for  Low-coefficient  Quartz  Crystals 


110 


is  very  stable,  is  little  affected  by  shocks,  and  ages  very  little  over  a  long  period  of  time. 

It  constitutes  an  oscillator  that  maintains  its  frequency  to  1  part  in  109  or  better  over  long 

periods  of  time  and  has  made  possible  the  precise  timing  necessary  in  the  Loran  system 

and  in  very  stable  time  standards  (see 

Spencer  Jones,  Endeavor,  Vol.  4,  No.  16 

[October  1945]).  X-CUT  CRYSTAL 

34.  PROPERTIES  OF 
ROCHELLE  SALT 

Rochelle  salt  is  sodium  potassium  tar- 
trate  with  four  molecules  of  water  of 
crystallization  (NaKC^Oe  •  4H2O)  and 
forms  in  the  orthorhombic  bisphenoidal 
class.  The  usual  form  of  the  crystal  is 
indicated  by  Fig,  9 (a),  which  shows  the 
directions  of  the  X t  Y,  and  Z  axes.  Since 
the  crystal  has  water  of  crystallization  it 
has  a  vapor  pressure.  As  shown  by  Fig.  ^)  45o_CUT 
10,  lower  line,  if  the  humidity  of  the  sur- 
rounding atmosphere  is  below  35  per  cent 
at  25  deg  cent,  the  water-vapor  pressure 
of  the  crystal  is  greater  than  the  vapor 
pressure  of  water  in  the  surrounding  at- 
mosphere and  the  crystal  will  lose  water 
and  dehydrate.  This  causes  a  white 
powder  of  dehydrated  material  to  form  on 
the  outside  of  the  crystal  which  will  ruin 
the  operation  of  the  crystal  if  it  becomes 
too  large.  The  crystal  is  stable  between 
35  and  85  per  cent  relative  humidity  (d)  TORSIONAL  X_M  (e)  5^  X.CUT 
Above  85  per  cent  humidity  the  crystal  Methods  for  Obtaining  Longitudinal,  Flex- 

will  absorb  water  from  the  atmosphere  on    urali  Torsional,  and  Pbue  Shear  Vibrations  from  an 
its  surface  and  will  slowly  dissolve  if  kept  X~cut  Rochelle  Salt  Crystal 


(C)  BENDER 


13-66 


ELECTROMECHANICAL-ACOUSTIC  DETICES 


In  such  an  atmosphere.  To  minimize  these  humidity  effects  the  crystals  are  often  coated 
with  waxes,  which,  however,  retard  rather  than  prevent  the  dehydration  of  the  crystal.  If 
the  crystal  can  be  hermetically  sealed  in  a  container  with  powdered  crystalline  Rochelle  salt 
and  dehydrated  Rochelle  salt,  it  can  be  made  to  last  indefinitely.  The  powdered  salt  will 
give  up  water  if  the  temperature  rises  and  the  dehydrated  salt  will  take  up  water  if  the 
temperature  lowers,  and  the  two  will  maintain  a  humidity  that  approximates  the  lower 
curve  as  a  function  of  temperature.  At  a  temperature  of  55  deg  cent  (130  deg  fahr)  the 
crystal  breaks  up  into  sodium  tartrate  and  potassium  tartrate  with  the  evolution  of  one 
mole  of  water  which  dissolves  the  two  crystals  in  a  liquid  solution.  If  this  solution  is 
rapidly  supercooled  it  remains  quite  fluid  for  a  number  of  minutes  before  it  crystallizes 
and  hardens.  This  "melted1'  Rochelle  salt  forms  a  very  stiff  glue  that  has  been  usedjjo 
glue  together  pieces  of  Rochelle  salt. 


VOLTS   PER 
CENTI  METER:  SOOf-J 
5 


O  X)  20  3O  40          5O 

TEMPERATURE   IN  DEGREES  CENTIGRADE 

FIG.  10.    Limits  of  Humidity  Stability  of  a 
Rochelle  Salt  Crystal 


-3O      -2O      -10          O          K)         2O        3O       40 
TEMPERATURE  IN   DEGREES  CENTIGRADE 


50 


FIG.  11.    Free  Dielectric  Constant  of  an  Z-cut  Rochelle 

Salt  Crystal  as  a  Function  of  Temperature  and  Field 

Strength 

Between  the  temperatures  of  —18  and  +24  deg  cent  Rochelle  salt  has  ferroelectric 
properties.  By  this  is  meant  that  it  becomes  spontaneously  polarized  in  the  ±x  direction. 
A  small  applied  field  causes  a  large  change  in  polarization  and  one  which  follows  a  hysteresis 
loop  as  does  a  ferromagnetic  material.  Since  the  piezoelectric  strain  is  proportional  to  the 
polarization,  a  large  distortion  of  the  crystal  occurs.  Hence  Rochelle  salt  is  principally 
used  when  a  large  motion  is  required  for  a  small  applied  voltage.  The  displacement,  how- 
ever, shows  a  hysteresis  effect  and  varies  considerably  with  temperature  for  a  given  applied 
voltage.  The  piezoelectric  equations  for  Rochelle  salt  can  be  written  in  the  form 
Si 


-f 


533^3 


(11) 


where  in  cgs  units  the  constants  have  the  values 

su  =  5.18  X  10"12    cm2  per  dyne        su?  =  7.98  X  10"12 

$n  =  3.49  X  KT12  sis0  =  32.8  X  10"12 


,  -  48  X  10~8 
=  -^  =  10.0 


=  3.34  X  IQ-& 


10.08  X  lO"1 


,  -  10.2 


«u 1.53  X  10~w  £U  =  62  X  10-* 

to  =  -2.11  X  10"12  £25  =  170  X  10~8 

$23  -  —1.03  X  10"12 

For  the  inks  system  the  elastic  compliances  are  multiplied  by  10,  the  piezoelectric  constant 
g  is  multipEed  by  S  X  10s,  and  the  dielectric  constants  are  multiplied  by  8.85  X  10"12. 


PROPERTIES   OF  ROCHELLE   SALT  13-67 

The  only  constant  in  the  above  equations  that  varies  widely  with  temperature  and  field 
strength  is  the  dielectric  constant  enT  (the  inverse  of  j3nT$.  For  low  field  strengths  and 
frequencies  above  1  kc,  the  dielectric  constant  as  a  function  of  temperature  is  shown  by 
the  dotted  line  of  Fig.  11.  It  rises  to  very  high  values  at  the  two  Curie  temperatures  —  18 
and  +24  deg  cent.  For  high  field  strengths  the  dielectric  constant,  as  shown  by  the  solid 
line  of  Fig.  11,  measured  by  the  average  slope  of  the  hysteresis  loop,  becomes  larger  be- 
tween the  Curie  points  than  it  is  at  the  Curie  temperatures. 

USEFUL  CUTS  IN  ROCHELLE  SALT.  The  cut  most  widely  used  is  the  X  cut,  which 
as  shown  by  Fig.  9(6)  is  cut  with  its  major  face  normal  to  the  X  axis.  If  a  voltage  is  applied 
to  this  cut  it  shears  so  that  the  square  changes  into  a  rhombus.  By  cutting  the  crystal 
length  45°  from  the  crystallographic  Y  and  Z  axis,  a  crystal  is  obtained  which  elongates 
along  one  direction  and  contracts  along  the  width.  This  cut  which  is  known  as  the  45° 
X  cut  is  widely  used  in  producing  longitudinal  vibrations.  By  combining  two  longitudinal 
crystals  as  shown  by  Fig.  9(c)  a  "bimorph"  crystal  is  obtained  which  bends.  This  has  a 
much  lower  frequency  than  a  longitudinal  crystal  and  is  used  in  voice-frequency  apparatus 
for  picking  up  and  reproducing  sound.  Figure  Q(d)  shows  a  combination  of  two  X-cut 
crystals  used  to  produce  a  twisting  motion.  The  center  faces  of  the  two  crystals  form  one 
set  of  electrodes  and  the  two  outside  electrodes  the  other  pair  so  that  two  opposing  face 
shears  are  applied  to  the  combination.  This  causes  the  whole  crystal  to  twist  and  produces 
a  torsional  motion  in  the  pair.  Finally  Fig.  9(e)  shows  two  thin  face  shear  X-eut  crystals 
which,  when  they  are  clamped  on  three  corners,  produce  a  large  motion  at  the  fourth 
corner.  All  three  of  these  bimorph  type-crystals  have  been  used  in  such  devices  as  phono- 
graph pickups,  microphones,  headphones,  loudspeakers,  surface-roughness  analyzers, 
and  light  valves  and  have  many  other  applications. 

For  a  45°  X-cut  crystal  the  equations  applicable  for  the  extension  are 


Si  - 

(12) 

Ex  =  -giTi  -f  &iT&x 

where  Si  is  the  strain  along  the  length,  TI  the  stress  applied  along  the  length,  gi  the  effective 
piezoelectric  constant  for  the  45°  axis,  and  @T  the  impermeability  (inverse  of  the  dielectric 
constant)  which  is  measured  when  the  crystal  is  free  to  move.  In  cgs  units  the  above 
constants  have  the  values 

aa'J)  :=  3.16  X  KT12    cms  per  dyne;        gi  =  31  X  10~s  -  ~  (13) 

while  the  free  dielectric  constant,  which  is  the  inverse  of  j5r,  has  the  value  shown  by  Fig.  11 
for  low  applied  fields  and  for  high  fields  (500  volts  per  centimeter).  Equations  (12)  can 
be  used  to  predict  the  action  of  the  crystal  under  static  conditions  or  at  frequencies  much 
lower  than  the  resonant  frequencies  of  the  crystal.  For  example,  if  we  wish  to  find  the 
response  of  the  crystal  as  a  microphone,  the  second  equation  states  that,  for  open-circuit 
conditions  for  which  the  charge  on  the  surface,  and  hence  the  electrical  displacement  Z>a, 
is  zero,  the  potential  generated  for  a  given  pressure  (negative  of  the  tension  TI)  is 

f 

Ex  =  —  ==  giTi  =  31  X  10"8     (pressure  in  dynes  per  cm*)  (14) 

It 

Since  the  electrostatic  unit  of  potential,  the  statvolt,  is  300  volts,  the  volts  generated  pear 
dyne  per  square  centimeter  pressure  are 

^voits  =  31  X  10~*  X  300  X  It  X  p  =  9-10  X  10~s     volt  per  dyne  per  sq.  cm.  for 

a  crystal  1  cm  thick     (15) 

Since  the  voltage  generated  for  a  given  pressure  is  directly  proportional  to  the  gi  constant, 
which  is  one-half  the  appropriate  shear  constant,  eqs.  (11)  shows  that  a  45°  F-cut  crystal, 
which  will  have  a  gi  piezoelectric  constant  equal  to  lfe  X  170  X  10~~8  =  85  X  10"~s,  will 
generate  about  3  times  the  open-circuit  voltage  for  the  same  pressure  that  a  45°  X-cut 
crystal  will.  The  45°  Y-cut  has  been  used  to  some  extent  as  a  microphone  and  as  a  trans- 
ducer in  underwater  sound  equipment  for  transforming  electrical  into  mechanical  energy. 
When  the  crystal  is  used  as  a  microphone  working  into  a  low  impedance,  the  F-ctit  will 
not  deliver  as  much  voltage  as  an  X-cut  crystal  on  account  of  the  very  low  impedance 
(high  capacity)  of  the  X-cut  crystal,  but  the  voltage  that  it  does  deliver  is  not  a  function 
of  the  temperature  as  is  the  voltage  of  the  45°  X-cut. 

By  eliminating  D*  from  eqs.  (12),  the  strain  Si,  which  is  the  expansion  per  unit  length, 
can  be  expressed  in  terms  of  the  applied  field  as 


13-68 


ELECTROMECHANICAL-ACOUSTIC  DEVICES 


In  the  absence  of  an  external  stress  Ti  the  total  free  displacement  of  1  volt  applied  is 

d  =  Sil  =  1.03  X  10-9  ^  E  ~  (17) 


This  displacement  as  a  function  of  the  volts  per  inch  applied  is  shown  by  Fig.  12  for  several 
different  temperatures.  Outside  the  Curie  region  the  displacement  is  much  less  since  the 
dielectric  constant  e^  is  so  much  smaller,  particularly  for  large  fields. 

When  two  crystals  are  glued  together  to  form  a  bimorph  unit  it  has  been  shown  (see 
W.  P.  Mason,  Electromechanical  Transducers  and  Wave  Filters,  p.  214,  Van  Nostrand)  that 
the  displacement  of  the  component  longitudinal  crystals  is  multiplied  by  the  factor 
3l/lt,  ^where  I  is  the  length  of  the  crystal  and  It  the  total  thickness  of  the  two  elements. 
This  is  a  method  of  enhancing  the  total  displacement  of  the  unit  at  the  expense  of  a  con- 
siderable lowering  of  the  resonant  frequency  of  the  device.  Since  the  dielectric  constants 
of  the  two  crystals  glued  together  will  be  less  than  the  free  dielectric  constant  of  Fig.  11 
and  will  approach  the  dielectric  constant  of  the  clamped  crystal  shown  by  Fig.  13,  the  very 


400 


£300 


X. 


VOLTS  PER 
CENTIMETER: 

•<—  500 


800 


50 


-3O      -2O       -IO          0  1O         20         3O       4O 

TEMPERATURE  IN  DEGREES  CENTIGRADE 

FIG.   12.     Clamped  Dielectric  Constant  of  an  X-cut 

Rochelle  Salt  Crystal  as  a  Function  of  Temperature  and 

Field  Strength 


0  200        400        60O        800      tOOO 

VOLTS  (PEAK)  PER  INCH   THICKNESS 

FIG.  13.    Strain  in  Rochelle  Salt  as  a  Func- 
tion of  Temperature  and  Field  Strength 


large  temperature  and  saturation  effects  noted  for  the  free  crystal  will  be  considerably 
reduced  for  the  bimorph  type.  However,  the  response  may  vary  by  a  factor  of  5  for  a 
wide  temperature  range.  A  typical  response  in  the  ferroelectric  range  for  a  bender  unit 
1  1/2  in.  long,  3/4  in.  ^ide,  and  0.040  in.  thick  is 


33  volts 

77  volts 

125  volts 

140  volts 


0.002  in. 
0.0045  in. 
0.006  in. 
0.0056  in. 


The  displacement  for  any  other  shape  unit  will  vary  in  proportion  to  the  factor  (I /It)2  and 
will  be  independent  of  the  width. 

When  such  units  are  used  as  voltage  generators  as  in  phonograph  pickup  devices,  the 
mechanicaMmpedance  of  the  device  is  very  considerably  lowered  over  what  would  be 
obtained  with  a  clamped  longitudinal  device.  The  response  can  be  calculated  by  deter- 
mining how  much  strain  is  generated  by  a  given  motion  and  calculating  the  voltage  from, 
eqs.  (12) .  A  typical  unit  0.030  in.  thick,  n/16  in.  iongj  ^d  7/16  in.  ^de  will  give  an  output 
as  high  as  1  volt  when  played  from  a  phonograph  record.  This  response  will  be  relatively 
independent  of  the  temperature  when  the  device  is  worked  into  the  grid  of  a  vacuum  tube. 


35.  PROPERTIES  OF  AMMONIUM  DIHYDROGEN 
PHOSPHATE  (ADP) 

Ammonium  dihydrogen  phosphate,  which  has  been  given  the  abbreviation  ADP,  is 
one  member  of  four  isomorphous  salts  whose  dielectric  properties  were  first  investigated 
by  Busch  (Neue  Seignette  Elektrica,  Heh.  Phys.  Acta,  Vol.  11,  No  3  [1938]).  Two  members 
of  this  group,  namely,  potassium  dihydrogen  phosphate  and  potassium  dihydrogen 
arsenate,  were  found  to  have  ferroelectric  effects  at  121  and  91  deg  absolute  temperature. 
Of  these  isomorphous  crystals  ADP  was  the  crystal  which  had  the  largest  piezoelectric 
coupling  (about  30  per  cent),  and  it  was  widely  used  during  World  War  II  as  the  trans- 
ducing element  for  underwater  sound  projectors  and  hydrophones.  It  appears  likely 
that,  for  devices  that  transform  mechanical  vibrations  into  electrical  vibrations — phono- 


AMMONIUM  DIHYDROGEN  PHOSPHATE   (ABP)        13-69 

graph  pickups,  microphones,  etc. — ADP  will  give  superior  results  to  Rochelle  salt  and 
may  eventually  replace  it  for  such  applications.  For  devices  that  have  to  produce  a  large 
motion  for  a  given  voltage,  however,  Rochelle  salt  is  still  the  only  crystal  that  has  a 
large  enough  du  constant  to  be  of  interest. 

ADP  crystallizes  in  the  tetragonal  scalenohedral  class  with  the  habit  shown  by  Fig.  14. 
The  c  or  Z  axis  lies  along  the  long  direction  of  the  crystal;  this  is  an  a*n's  of  fourfold  alter- 
nating symmetry.  The  X  and  7  axes  lie  normal  to 
the  prism  faces;  they  are  axes  of  two-fold  symmetry. 
Since  the  properties  of  crystals  cut  normal  to  these 
two  surfaces  are  identical  it  is  a  matter  of  conven- 
tion which  is  called  X  and  which  7.  The  two 
diagonal  axes,  labeled  PI  and  Pzt  can  be  distinguished 
by  piezoelectric  tests,  and  PI  has  been  taken  as  that 
axis  along  which  a  positive  stress  (tension)  produces 
a  positive  charge  at  the  positive  (i.e.,  the  upper)  end 
of  the  Z  axis.  With  the  Z  axis  vertical  and  the  PI 
axis  toward  the  observer's  right  hand,  the  X  axis 
has  been  taken  as  the  axis  that  runs  from  front  to 
back  of  the  crystal  and  the  7  axis  the  one  that  runs 
from  left  to  right. 

ADP,  which  has  the  chemical  formula  NH-iHsPO^ 
has  no  water  of  crystallization  and  hence  will  not 
dehydrate  when  the  humidity  becomes  low.  At 
about  93  per  cent  humidity,  the  crystal  will  del- 
iquesce and  will  pick  up  water  from  the  atmos- 
phere. In  practice  it  is  necessary  to  keep  the  crys- 
tal in  an  atmosphere  for  which  the  humidity  is  50 
per  cent  or  less  since  the  water  collected  on  the  pt 
surface  provides  a  leakage  path  across  the  crystal 
edges  which  becomes  low  enough  to  cause  trouble 
for  humidities  above  50  per  cent.  Owing  to  the  hy- 
drogen bond  system  ADP  also  has  a  volume  leakage 
which  for  a  pure  salt  is  shown  by  Fig.  15.  For  the 
Z-cut  crystal  having  a  dielectric  constant  of  15.7 
this  leakage  will  impair  the  response  only  for  fre- 
quencies below  1  cycle  per  second.  However,  cer- 
tain impurities  introduced  by  the  growing  process 
can  markedly  decrease  this  resistivity,  and  for  some 
applications  it  is  necessary  to  specify  a  high  resistivity.  ADP  can  be  taken  up  to  ISO  deg 
cent  before  it  melts.  However,  ammonia  is  given  off  from  the  surface  at  temperatures 
above  100  deg  cent,  and  since  this  impairs  the  adherence  of  the  electrodes  to  the  crystal 
surface  it  is  desirable  to  keep  the  temperature  of  operation  under  100  deg  cent.  The 
crystal,  therefore,  is  useful  under  any  likely  ambient  temperature  conditions. 

The  piezoelectric  equations  for  ADP  take  the  form 


FIG.   14.     45°   Z-cut  ADP   Crystal   and 
Form  of  Natural  Crystal 


£3 


4- 


(18) 


S* 

where  in  cgs  units  the  constants  have  the  following  values: 

«u  =  1.74  X  10~12  cm2  per  dyne        SUD  =  11.4  X  KT12  cm2  per  dyne        *?  =  59.0 
Sl2  =  0.7  X  10-12  s*P  =  14.7  X  10-13  ef  =  15.7 

sn  -  - 1.1  X  10"12  £14  =  1-06  X  10-*  <19) 

533  =  4.35  X  10"12  gx>  =  HS.5  X  10~8 

TTSEFtTL  CUTS  FOR  ADP  CRYSTALS.  Since  the  gas  constant  is  so  much  larger  than 
the  £14  constant  in  ADP  it  is  obvious  that  most  of  the  useful  cuts  will  be  those  that  are 
normal  or  nearly  normal  to  the  Z  axis.  A  crystal  cut  normal  to  the  Z  axis  will  generate  a 
face  shear  motion  similar  to  that  shown  by  Fig.  9(6)  for  Rochelle  salt.  Hence  by  cutting 


13-70 


ELECTROMECHANICAL-ACOUSTIC  DEVICES 


the  length  45°  from  the  X  and  F  crystallographic  axes  a  longitudinal  motion  may  be  pro- 
duced. The  Z  and  the  45°  Z  cut  are  the  principal  ones  used  for  ADP  crystals.  The  Z  cut 
has  been'used  in'generating  face  shearing  modes  and  in  the  production  of  torsional  crystals. 


!0IOXK> 
8 
6 


TEMPERATURE  IN   DEGREES  CENTIGRADE 
9O    8O     70      6O    50      4O        3O       20         10  O 


xto3 


Z 

>     0-2 

i 
i2   o,t 

O.O6 
0.04 


2  CUT  KDP 


4- 


fr 


Z  CUT  ADP 


£45 


I 


30 


270 


310 


350 


370 


O  2  4.  6  S 

THICKNESS  IN  MILLIMETERS 


FIG.  15.    Resistivity  of  ADP  and  KDP  as  a  Function  of       FIG.    16.     Breakdown   Voltage   for   an 
Temperature  ADP  Crystal  as  a  Function  of  Thickness 

The  45°  #-cut  crystal  has  been  used  as  the  transducing  element  in  underwater  sound  equip- 
ment and  in  microphones,  in  phonograph  pickups,  and  in  devices  for  transforming  mechan- 
ical energy  into  electrical  energy. 

The  equations  of  motion  of  a  45°  Z  cut  take  the  form 

',  +  flg  (20) 


Si  = 


where  gi  =  gu/2  =  59.2  X  10~8;  s^P  =  4.72  X  1CT12  cm2  per  dyne; 

*-&-  15J 

When  a  crystal  1  cm  thick  is  used  as  a  voltage-generating  device  the  number  of  volts  gen- 
erated on  open  circuit  per  dyne  per  square  centimeter  is 

-Evoits  =  1.78  X  1<T4    volt  (21) 

This  is  larger  than  for  45°  X-cut  Rochelle  salt.  On  account  of  the  lower  dielectric  constant 
this  crystal  has  to  be  worked  into  a  higher  impedance  than  Rochelle  salt  to  obtain  the 
same  output.  Crystals  of  this  sort  are  replacing  Rochelle  salt  for  applications  such  as 
microphones  and  phonograph  pickups  on  account  of  their  greater  chemical  stability  and 
their  ability  to  withstand  wide  temperature  variations. 

The  electromechanical  coupling  factor  of  ADP  is  given  by  the  formula 


AMMONIUM  DIHYDROGEN  PHOSPHATE   (AD?)        13-71 


As  can  be  seen  from  eq.  (9)  these  crystals  can  convert  electrical  into  mechanical  energy, 
or  vice  versa,  efficiently  over  a  frequency  range  of 


7 


(23) 


Considerable  amounts  of  power  can  be  transformed  from  electrical  into  high  mechanical 
impedance  systems.  The  crystal  limitations  are  the  breaking  strain  and  the  voltage 
gradient  that  the  crystals  will  stand.  Experiments  with  ADP  crystals  show  that  they 
will  break  if  the  strain  exceeds  from  4  to  10  X  10~*  cm  per  cm.  The  voltage  gradient 
that  they  will  stand  before  a  voltage  puncture  occurs  is  a  function  of  the  thickness  of  the 
crystal.  Figure  16  shows  the  voltage  gradient  that  will  produce  a  puncture  on  the  average 
crystal.  It  can  be  shown  that  the  particle  velocity  on  the  end  of  a  quarter-wavelength  or 
half-wavelength  crystal  is  equal  to 


|  =  US;*     or    J  -  3.3  X  10*Sjf  (24) 

where  f  is  the  particle  velocityt  t>  is  the  velocity  of  propagation,  and  SM  is  the  maximum 
strain  that  the  crystal  will  suffer.  This  maximum  strain  occurs  at  the  middle  of  a  half- 
wavelength  unit  or  at  the  glued  joint  of  a  quarter-wavelength  unit.  Since  the  crystal  is 
stronger  than  most  of  the  adhesives  that  can  be  used  to  attach  it  to  high-mechanical- 
impedance  solid  materials,  the  half-wavelength  crystal  can  be  used  to  produce  more  power 
output  than  the  quarter-wavelength  unit.  The  two  types  of  units  are  shown  by  Fig.  17. 


RADIATING 
FACE 


METAL 

QUARTER-WAVE 
BACKING    PLATE 


QUARTER -WAVE 
CRYSTAL 


LOW  MECHANICAL 

IMPEDANCE 


(3)  QUARTER-WAVE  TRANSDUCER  (t)")   HALF-WAVE   TRANSDUCER 

FIG.  17.     Quarter-  and  Half-wave  Transducers 

The  quarter-wavelength  unit  is  glued  to  a  heavy  metal  backing  plate  and  radiates  its 
energy  from  its  free  face.  A  half-wavelength  unit,  on  the  other  hand,  works  into  a  low 
mechanical  impedance  on  one  end  and  radiates  its  energy  from  the  other  end.  For  a 
half-wavelength  unit  the  maximum  strain  that  the  crystal  can  safely  stand  is  4  X  10"*,  so 
that  the  maximum  particle  velocity  obtainable  for  an  ADP  crystal  is  about 


132    cm  per  sec 


(25) 


If  the  particle  velocity  is  working  into  a  high  mechanical  impedance  such  as  the  radiation 
impedance  of  water,  which  is  R  X  1.5  X  10s  mechanical  ohms  per  square  centimeter,  the 
energy  radiated  with  this  particle  velocity  is 

£?(CTg8/sec/sQ/cm>  =  lm2-R  —  2.6  X  10s     ergs  per  sec  per  sq  cm  (26) 

=  260    watts  per  sq  cm 

This  power  usually  exceeds  the  power  allowed  by  the  voltage  puncture  limit  (unless  the 
crystals  are  made  very  thin)  and  this  is  the  usual  crystal  limitation. 

It  can  be  shown  that  for  a  half-wavelength  ADP  crystal  radiating  into  a  mechanical 
load  of  RM  mechanical  ohms  per  square  centimeter  (ratio  of  force  in  dynes  per  square 
centimeter  to  velocity  in  centimeters  per  second)  the  power  radiated  in  watts  per  square 
centimeter  is  given  in.  terms  of  the  volt-age  gradient  in  volts  per  centimeter  and  the  mechan- 
ical load  jRjw  by  the  equation 


Power  - 


(half-wavelength  radiator) 


(27) 


For  a  quarter-wavelength  unit,  the  power  radiated  is  one-fourth  of  this  for  the  same  voltage 
gradient  or 

Power  =   ^J  (28) 


13-72  ELECTROMECHANICAL-ACOUSTIC  DEVICES 


Hence  if  we  know  the  limiting  voltage  gradients,  the  amount  of  power  that  the  crystal 
will  radiate  before  it  punctures  can  be  calculated.  For  example,  if  a  half-wavelength 
radiator  is  working  into  the  radiation  impedance  of  water  and  is  made  up  of  crystals  0.25 
cm  thick,  the  crystals  should  withstand  a  voltage  gradient  of  50,000  rms  volts  per  centi- 
meter. From  eq.  (27)  the  power  radiated  should  be  660  watts  per  square  centimeter 
before  dielectric  breakdown  occurs.  Crystal  fracture  then  becomes  the  limiting  factor. 
In  practice  the  limitation  of  power  does  not  lie  with  the  crystal  but  occurs  in  the  medium 
if  this  is  liquid  or  in  the  glued  joint  if  a  solid  is  used  to  transmit  the  power.  A  usual  figure 
for  continuous  power  is  5  watts  per  square  centimeter.  If  the  crystal  is  to  radiate  into  a 
gas  such  as  air,  the  limiting  power  is  invariably  determined  by  the  limiting  strain  that  the 
crystal  can  stand  before  it  breaks.  From  eq.  (28)  the  maximum  amount  of  power  that 
can  be  radiated  into  air  by  an  ADP  crystal  is  0.075  watt  per  square  centimeter  since  the 
radiation  impedance  of  air  is  43  ohms  per  square  centimeter.  A  crystal  is  not  an  efficient 
means  for  exciting  an  air  vibration. 

Since  the  limiting  particle  velocity  for  an  ADP  crystal  is  about  130  cm  per  sec,  a  crystal 
cannot  be  used  directly  to  produce  very  high  strains  in  metals  or  terminal  velocities 
approaching  the  speed  of  sound.  If,  however,  a  crystal  mosaic  is  glued  to  a  metal  rod 
tapered  exponentially  like  a  horn,  a  very  high  strain  and  a  very  high  terminal  velocity  can 
be  produced  at  the  small  end.  Figure  18  shows  a  construction  proposed  by  the  writer  for 


Y  WAVELENGTH 
(WHERE  n=JNTEGER) 


HALF-WAVE  CRYSTAL 


FIG.  18.     Mechanical  Horn  for  Producing  a  Large  Strain  in  a  Metal  Sample 


testing  fatigue  in  metals.  A  crystal  mosaic  several  inches  in  cross-section  is  glued  to  a 
steel  rod  which  tapers  from  the  crystal  area  down  to  a  thickness  of  0.05  in.,  after  which  it 
increases  in  diameter.  The  taper  is  an  exponential  function  of  the  length  and  must 
satisfy  the  relation 

T  £  ^  (29) 

where  the  taper  T  is  determined  by  the  equation  for  the  area 

S 


(30) 

where  /  is  the  resonant  frequency  of  the  crystal,  and  v9  the  velocity  of  sound  in  the  steel. 
If  the  total  length  of  the  steel  piece  is  made  an  integral  number  of  half  wavelengths  of 
the  frequency,  the  glued  joint  will  come  at  a  loop  of  the  motion  and  will  not  be  appreciably 
strained.  The  whole  system  will  act  as  a  resonant  system  and  will  produce  a  considerable 
motion  for  small  applied  voltages.  The  steel  section  adjacent  to  the  crystal  will  have 
the  same  particle  velocity  as  the  crystal  surface.  Now  it  can  be  shown  that  the  strain  in 
the  bar  of  uniform  section  at  the  nodal  point  (point  of  maximum  strain)  is  equal  to 

S  -  f-  (31) 

vt 

where  v9  is  the  velocity  of  propagation  of  the  wave  in  the  steel  and  £  the  particle  velocity 
at  the  surface.  The  effect  of  the  tapered  section  is  to  increase  the  particle  velocity  in 
inverse  proportion  to  the  diameter.  Hence  if  the  diameter  decreases  from  2  in.  to  0.05  in. 
the  velocity  is  multiplied  by  a  factor  of  40  and  the  strain  at  the  nodal  point  is  equal  to 


S.  =  40  X  -  S 


(32) 


where  Sx  is  the  strain  in  the  steel,  Sc  the  strain  in  the  crystal,  ve  the  velocity  of  propagation 
in  the  crystal  (3.3  X  105  cm  per  sec),  and  va  the  velocity  of  propagation  in  the  ste^l  (about 


AMMONIUM  DIHYDROGEN  PHOSPHATE    (ADP)         13-73 

5.1  X  10s  cm  per  sec).  Hence,  for  a  strain  of  4  X  10~4  in  the  crystal,  a  strain  of  0.01  can 
be  generated  in  the  steel.  This  is  sufficient  to  cause  plastic  deformation  in  the  steel,  and 
by  gradually  increasing  the  drive  on  the  crystal  the  fatigue  properties  of  the  steel  can  be 
investigated  at  a  high  rate  of  strain  and  of  velocity. 

This  same  system  can  be  used  to  produce  a  high  particle  velocity  on  the  small  end  of 
the  steel  bar.  The  only  limitation  is  the  strain  that  the  metal  will  stand.  Other  uses 
appear  to  be  delivering  a  large  amount  of  power  for  a  small  area.  By  mounting  a  torsional 
crystal  on  the  large  end  of  the  bar  a  torsional  vibration  can  be  given  to  the  bar  and  the 
properties  of  the  material  under  shearing  strain  can  be  tested.  An  ADP  crystal  can  be 
made  to  vibrate  in  torsion  by  using  the  electrode  system  shown  by  Fig.  19.  The  inside 


(a)  (b) 

FIG.  19.     Method  for  Cutting  an  ADP  Crystal  to  Obtain  a  Torsional  Oscillation 

surface  is  covered  by  one  electrode  while  the  two  outside  electrodes,  each  of  which  covers  a 
90°  segment,  are  connected  together  and  form  the  other  electrode.  The  centers  of  the 
two  outside  electrodes  are  normal  to  the  Z  axis,  and  the  field  is  directed  out  from  the 
center  for  both  electrodes  as  shown  by  Fig.  19(6),  thus  producing  a  shearing  motion  for 
one  segment  and  the  opposite  shear  on  the  other  segment  so  that  the  whole  crystal  is 
given  a  torsional  motion. 


SECTION  14 
OPTICS 


GEOMETRICAL  OPTICS 
ABT.  BY  D.  W.  EPSTEIN  PAGE 

1.  Reflection  and  Refraction 02 

2.  Lenses 09 

3.  Photometry 14 

4.  Light  Measurement 17 

5.  Photometric  Relations  in  Non-visual  Op- 

tical Systems IB 

6.  Reflective  Optical  System  for  Television 

Projection 20 

VISION 

BY  KENNETH  N.  OGLE 

7.  The  Structure  of  the  Eye 25 

8.  The  Optical  Characteristics  of  the  Eye.     29 


AST.  PAGE 

9.  The  Light  Sense  ..............  .  .....  .  30 

10.  Temporal  Aspects  of  Perception  .......  33 

11.  Color  .................  .  .............  35 

12.  The  Space  Sense  ----  .  ................  39 

13.  Binocular  Vision  .....................  46 


ELECTRON  OPTICS 
BY  D.  W.  EPSTEIN 

14.  Electrostatic  Lenses  ..................     51 

15.  Magnetosiatic  Leases  ................     59 

16.  Electron  Prisms  .....................     62 

17.  General  Theorems  on  Electron  Optical 

Systems  ..........................     63 


144)1 


OPTICS 
GEOMETRICAL  OPTICS 

By  B.  W.  Epstein 

Geometrical  optics  is  mainly  concerned  with  the  geometrical  relations  of  the  propaga- 
tion of  radiant  energy. 

Neglecting  quantum  effects,  the  propagation  of  radiant  energy  is  governed  by  Maxwell's 
equations.  Fermat's  principle  of  least  time,  which  is  the  fundamental  law  governing  the 
propagation  of  "rays"  in  geometrical  optics,  follows  from  Maxwell's  equations  if  the 
wavelength  of  the  radiation  is  allowed  to  approach  zero.  Although  geometrical  optics 
applies  strictly  only  to  the  propagation  of  radiation  of  zero  wavelength,  it  provides  a  very 
good  and  extremely  useful  approximation  to  any  case  where  the  wavelength  is  negligibly 
small  in  comparison  with  the  smallest  linear  dimension  of  the  apparatus. 

Fermat's  principle  of  least  time  states  that  the  path  traversed  by  light  in  passing  be- 
tween two  points  is  that  which  will  take  the  least  time.  The  general  law  expressed  by 
Fermat's  principle  is  also  known  as  the  law  of  extreme  path.  It  is  stated  mathematically  as 


0,  (1) 

_y  =  —  =  index  of  refraction. 

C  =  velocity  of  radiation  in  vacuum. 
V  =  velocity  of  radiation  in  medium. 
dS  =  element  of  path  length. 

The  product  of  index  of  refraction  and  path  length,  2V  dS,  is  known  as  the  optical  length 
of  a  "ray  of  light"  or  optical  distance.  Equation  (1)  states  that  a  light  ray  going  from 
point  A  to  point  B  will  always  choose  that  path  which  will  make  the  optical  distance  an 
extremum  (generally  a  minimum  but  sometimes  a  maximum)  with  respect  to  all  neigh- 
boring paths  for  rays  of  the  same  frequency.  The  laws  of  linear  propagation,  reflection, 
and  refraction  may  be  deduced  from  Fermat's  principle. 

Although  the  frequency  is  a  constant  for  a  given  radiation,  and  its  wavelength  varies 
with  the  medium  traversed,  it  has  become  customary  to  specify  radiation,  especially 
visible  radiation,  by  its  wavelength  in  vacuum.  This  is  due  to  the  fact  that  fundamental 
measurements  yield  wavelength  rather  than  frequency.  The  units  of  wavelength  com- 
monly used  hi  optics  are: 

micron  (ju)  =  10~6  meter 

millimicron  (mju)  =  10""5  meter 

angstrom  unit  (A)  =  10  ~~10  meter 

The  visible  spectrum  extends  from  about  0.39  to  0.75  ju;  or  390  to  750  mp.;  or  3900  to 
7500  A;  or  from  about  4.0  X  1C14  to  7.7  X  1014  cycles  per  second. 

1.  REFLECTION  AND  REFRACTION 

When  a  beam  of  radiant  flux  or  luminous  flux  strikes  a  boundary  separating  two  homo- 
geneous isotropic  media,  it  is  in  general  partly  reflected  and  partly  refracted.  If  the  bound- 
ary is  smooth  (relative  to  the  wavelength  of  radiation),  the  following  simple  laws  of  refrac- 
tion and  reflection  apply  (see  Fig.  1) . 

LAW  OF  REFRACTION  OR  SWELL'S  LAW.  The  ratio  of  the  sine  of  the  angle  of 
incidence  to  the  sine  of  the  angle  of  refraction  is  constant  depending  only  on  the  indices  of 
refraction  of  the  two  media;  i.e., 

ATi  sin  Ji  =  AT2  sin  /*  (2) 

The  incident  ray,  the  refracted  ray,  and  the  normal  to  the  surface  at  the  point  of  incidence 
all  lie  in  the  same  plane. 

14-02 


REFLECTION  AND  REFRACTION 

z  a  2 


14-03 


PB*=  REFRACTED  RAY 
12    =  ANGLE  OF  REFRACTION 
Nt    =   INDEX  OF  REFRACTION  OF 
UPPER  MEDIUM 

N£    =  INDEX  OF  REFRACTION  OF 
LOWER  MEDIUM 


PQ  =  BOUNDARY  OF  TWO  MEDIA 

P    -  POINT  OF  INCIDENCE 

AP=  INCIDENT  RAY 

ZP=  NORMAL  TO  SURFACE  PQATP 

PB=  REFLECTED  RAY 

II  =  ANGLE  OF  INCIDENCE 

lz  '  ANGLE  OF  REFLECTION 

CoJ  O) 

FIG.  1.    Reflection  and  Refraction  of  Light  Ray 

LAW  OF  REFLECTION.    The  angle  of  reflection  is  equal  to  the  angle  of  incidence;  i.e., 

I*  =  -Ii  (3) 

The  incident  ray,  the  reflected  ray,  and  the  normal  to  the  surface  at  the  point  of  incidence 
all  lie  in  the  same  plane. 

ijOOr 


350      400       450       500       550       600      650      700 
\  IN  M>1 

FIG.  2.    Reflectance  of  Some  Metals  as  a  Function  of  Wavelength 

Reflection  from  smooth  surfaces  is  called  specidar  or  regular  reflection.  If  the  boundary 
is  between  a  transparent  dielectric  (air,  glass)  and  a  metal,  then,  in  general,  most  of  the 
incident  flux  is  reflected.  Figure  2  gives  the  specular  reflectance  (ratio  of  reflected  to 
incident  flux)  of  various  metals  as  a  function  of  wavelength. 


14-04 


OPTICS 


The  specular  reflectance  r/  at  tl^e  boundary  of  two  transparent  dielectrics  is  specified 
for  normal  incidence  by  Fresnel's  equation: 


rf 


(AT2  - 


A~i)2 


(4) 


where  JVi  and  AT2  are  the  indices  of  refraction  of  the  two  media. 

Figure  3  shows  the  reflectance  for  unpolarized  light  as  a  function  of  angle  of  incidence 

for  light  traveling  from  a  lower-index  medium  into  a  higher-index  medium  (N%  >  NI). 

It  is  seen  that,  as  the  angle  of  incidence  is 
N2>N»  increased,  the  reflectance  rises  gradually 

at  first,  and  then  rapidly,  until  it  becomes 
unity  at  the  angle  of  incidence  of  90°. 
The  remainder  of  the  incident  flux  enters 
the  medium  of  index  N%  at  the  angle  of 
refraction  /2  given  by  SnelTs  law,  eq.  (2). 
The  Fresnel  reflection  can  be  greatly 
reduced  by  means  of  a  transition  layer 
between  the  two  transparent  dielectrics. 
For  normal  incidence  and  for  a  particular 
wavelength,  the  Fresnel  reflection  can  be 
reduced  to  zero  if,  as  in  transmission  lines, 
the  index  of  refraction  of  the  transition 
layer  is  N  —  vNiNz  and  the  thickness  of 
the  layer  is  X/4. 

For  light  traveling  from  a  higher-index 
medium  into  a  lower-index  medium 
(Nz  <  JVi),  a  similar  behavior  is  observed 
up  to  a  certain  critical  angle  of  incidence, 
the  critical  angle  being  determined  by  the 
condition  that 


10*    20     30*    40*     50*    60*    70*    80" 


ANGLE  OF  INCIDENCE  IN  MEDIUM 


since  at  this  angle  of  incidence  the  angle 
of  refraction  is  90°  (sin  72  =  1).  For 
angles  of  incidence  greater  than  the  criti- 
cal angle,  the  beam  is  totally  reflected 
into  the  initial  medium.  Table  1  gives 


JIG.  3.    Fresnel  Reflectance  of  Unpokrized  Light  as  a   the  index  of  refraction  and  critical  angle 

of  incidence  relative  to  air  for  some  solids 
NI  into  Medium  of  Higher  Index  #2  and  liquids. 

Table  1.    Index  of  Refraction  and  Critical  Angle  of  Substances  Relative  to  Air 


Solids 


Substance 

tfD 

Ic 

Substance 

ND 

Ic 

Magnesium  fluoride  

.38 

46°  26' 

Water      

.334 

4  8o  33' 

Quartz  (fused)  

.458 

43°  18' 

Ether  

.357 

47°  28' 

Pyrex  (glass)  

.474 

42°  43' 

Alcohol  (ethyl) 

364 

47°  9' 

Methyl  methacrylate  

,49 

42°  9' 

Glycerine                ... 

471 

42°  50' 

Potassium  chloride  

.49 

42°  9' 

Carbon  bisulfide  

.630 

37°  51' 

Canada  balsam  

526 

40°  57' 

Methylene  iodide 

732 

35°  16' 

Sodium  chloride   .  .    . 

544 

40°  22' 

50%  Methylene  iodide  &  50% 

Polystyrene  

59 

38°  58' 

phosphorus 

1   929 

31°  14' 

Willemite 

71 

35°  47' 

Calcium  tungstate  ....... 

92 

31°  23' 

Zinc  oxide  

2  05 

29°  12' 

Zinc  $u]firip.  ,  n  ,  T  T  ,..,,,  x 

2.37 

24°  57' 

rHftTTWTwl  .    T  T  ,,  t  ......  x   .,.  v 

2  417 

24°  26' 

Cadmium  sulfide  

2.52 

23°  23' 

Liquids,  1 5  deg  cent 


DIFFUSE  REFLECTION,  If  the  boundary  between  two  media  is  rough,  specular  re- 
flection can  be  considered  to  exist  at  a  great  number  of  small  smooth  areas  oriented  in 
various  directions,  and  the  reflected  energy  is  distributed  over  a  wide  range  of  angles.  In 
general,  practical  surfaces  (boundaries)  reflect  partly  specularly  and  partly  diffusely. 


REFLECTION  AND  REFRACTION 


Figure  4  illustrates  this.  Pure  specular  reflection  is  shown  in  (a) ;  part  specular  and  part 
diffuse,  such  as  would  occur  at  most  matte  surfaces,  is  shown  in  (6) ;  pure  or  ideal  diffuse 
reflection  is  shown  in  (c).  In  perfectly  diffuse  reflection,  the  flux  reflected  per  unit  solid 
angle  is  proportional  to  the  cosine  of  the  angle  measured  from  the  normal  to  the  surface. 
This  statement  is  known 
as  Lambert's  law.  2 

DISPERSION.  The 
variation  of  the  refractive 
index  of  a  substance  with 
the  wavelength  (color)  of 
the  transmitted  light  is 
termed  dispersion.  The 
index  of  refraction  of  most 
glasses  varies  with  wave- 
length in  a  manner  which 
may  be  approximated  by 
the  dispersion  formula  of 
Cauchy: 


V 


PURE  SPECULAR  REFLECTION 
(cO 


PART  SPECULAR  AND  PART 
DIFFUSE  REFLECTION  DISTRIBU- 
TION A  FUNCTION  OF  I, 


IDEAL  DIFFUSE  REFLECTION 
DISTRIBUTION  INDEPENDEKT  OF  It 
(C) 

FIG.  4.    Specular  and  Diffuse  Reflection 


The  refractive  index  of 
optical  glass  is  generally 
measured  with  certain  defi- 
nite wavelengths  or  lines 
of  the  spectrum.  It  has 
become  customary  to  use 
the  spectral  lines  A' 
(0.7665  ju) ;  C  (0.6563  M)  ;  D  (0.5893  /*);  F  (0.4S61  AC) ;  G'  (0.4341  /*).  The  differences  in 
refractive  index  JVb  —  NC,  N-p  —  JVb,  NG'  —  JVp,  JVb  —  N&.',  and  Ny  —  NQ  are  taken 
as  a  measure  of  the  dispersion  of  the  glass  in  the  different  parts  of  the  spectrum.  If  a 
glass  is  specified  by  only  one  index  of  refraction  JVb  is  generally  meant.  A  quantity 
related  to  dispersion  which  is  in  very  general  use  utilizes  the  F  (blue),  D  (yellow),  and 
C  (red)  portions  of  the  spectrum  and  is  designated  by: 


-  1 


(6) 


The  V  values,  NA'»  NC,  ND,  7V>,  and  NQ'f  for  some  Bausch  &  Lomb  Optical  Company 
glasses  are  given  in  Table  2. 

Table  2.    Indices  of  Refraction  and  F-ntunber  of  Some  Optical  Glasses  Made  by  Bausch 

and  Lomb  Optical  Co. 


NA' 

Nc 

NT> 

N? 

NG' 

tfD-l 

Type  of  Glass 

766.5  mn 

656.3  rnjii 

589.3  m^ 

486.  1  mji 

434.1  HIM 

r        NY  -  Nc 

Borosilicate  Crown  BSC-I  .  .  . 
Crown  G-l  

1.50578 
1.51729 

.50860 
.52036 

1.51100 
1.5SSOO 

.51665 

.  52929 

1.52114 
1.53435 

63.5 
58.6 

Light  Barium  Crown  LBC-1  . 
Dense  Barium  Crown  DBC-1  . 
Crown  Flint  CF-l  

1.53529 
1.60439 
1.52217 

.53842 
.60793 
.52560 

1,54110 
1.61100 
1.5S860 

.54746 
.61832 
.53584 

.55257 
.62421 
.54178 

59.9 
58.8 
51.6 

Light  Barium  Flint  LBF-  1  ... 
Extra  Light  Flint  ELF-1  
"Barium  Flint  BF-2  -  - 

1.58110 
1.55086 
1   59682 

.58479 
,55495 
.60130 

1.58800 
1.55850 
1.60550 

.59580 
.56722 
.61518 

.60212 
.57447 
.  62345 

53.4 
45.5 
43.6 

Light  Flint  LF-I  

1.56425 

.56861 

1.57250 

.  58208 

.59011 

42.5 

Dense  Flint  DF-2  ... 

1  .  60684 

.61218 

1.61700 

.62904 

.  63929 

36.6 

Dense  Barium  Flint  DBF-  1  .  . 
Extra  Dense  Flint  EDF-3.  .  . 

1.60731 
1.70555 

.61242 
.71309 

1.61700 
1.7SOOO 

.62843 
.73766 

.63811 
.75304 

38.5 
29.3 

REFRACTION  AJTD  REFLECTION  AT  SPHERICAL  SURFACES.  Because  of  the 
relative  ease  of  manufacture,  most  optical  systems  consist  of  a  series  of  spherical  surfaces, 
a  plane  being  the  limiting  case  of  a  spherical  surface  of  infinite  radius  of  curvature.  By 
determining  the  plane  containing  the  ray  to  be  traced  and  the  center  of  curvature,  any 
problem  of  refraction  or  reflection  at  a  spherical  surface  may  be  reduced  to  a  problem  in 
plane  trigonometry. 


14-06 


OPTICS 


SIGN  CONVENTION.  Referring  to  Fig.  5,  let  the  paper  represent  the  plane  of  inci- 
dence containing  A\Q,  the  ray  to  be  traced  through  the  refracting  surface,  and  C  the  center 
of  curvature  of  the  spherical  surface.  The  signs  of  quantities  in  optical  calculations  have 
not  been  standardized,  but  the  sign  convention  indicated  in  Fig.  5  is  very  widely  used. 


FIG.  5.     Refraction  at  a  Spherical  Surface 

Distances  measured  to  the  right  of  the  pole  A  (or  center  of  curvature  C)  are  positive; 
those  to  the  left  of  the  pole  A  are  negative.  Distances  above  the  axis  are  positive,  and 
those  below  the  axis  are  negative.  Angles  shown  clockwise  are  positive;  those  shown 
counterclockwise  are  negative. 

EXACT  RAY  TRACING  EQUATIONS.  Using  the  above  sign  convention,  the  stand- 
ard ray  tracing  equations  given  below  follow  from  the  law  of  refraction  and  plane  trig- 
onometry. It  is  assumed  that  the  quantities  NI,  Nz,  R,  and  Si  (or  Pi)  are  given  and  that 
Ui  is  assigned  an  arbitrary  value  for  each  ray.  The  problem  is  to  find  7i,  lz,  C/2,  and  82 
(or  P2). 

sin  /i  =  — ^5 —  sin  U\  —  -~  sin  Ui  (7) 


R 


— 


-    r  *   •    r 

sin  Ja  =  -r=-  sin  Ji 

N2 


Uz  =  Ui  +  I] 
,  -  R  =  P2  =  R 


.-Is 

sin  Iz 
sin  Uz 


(8) 

(9) 

(10) 


In  dealing  with  a  complete  optical  system  with  many  surfaces  the  results  of  one  surface 
are  taken  as  the  initial  data  for  the  next  surface.  The  following  equations  which  may  also 
be  derived  with  the  aid  of  Fig.  5  have  been  found  very  useful  in  optical  calculations: 

(11) 
(12) 
(13) 


Vz  sin  272  —  Pi-ZVi  sin  I 
UQ  =•  Ui  +  Ii  - 
h  =  R  sin  UQ 
i  cos  Ii       Nz  cos  Iz      NZ  cos  Iz  —  NI  cos  I. 


+  Ia 


R 


-  COS  UQ 


(14) 


Since  at  reflection  Iz  =  —  Ii,  sin  I*  =  —  sin  /i,  the  law  of  reflection  may  be  treated  math- 
ematically as  a  particular  ease  of  the  law  of  refraction,  i.e.,  where  AT2  =  —  JVi.  Hence  the 
refraction  formulas  given  above  apply  to  the  case  of  reflection  at  a  spherical  surface  by 
simply  letting  N$  =  —JVi.  Thus  for  a  spherical  mirror,  the  above  equations  become 

(Fig.  6): 

•p 

sin  /i  =  — 1  sin  Ui  (15) 


sin  Is  =  —  sin  Ji 


P2sin  Uz 
_1  1 
Pi  Pi 


sin  t/2 
-Pi  sin  Ui 
2  cos  Ua 


(16) 
(17) 

(18) 
(19) 
(20) 


REFLECTION  AND  REFRACTION 


14-07 


FIG.  6.     Reflection  at  a  Spherical  Surface 


PARAXIAL  FORMULAS.  The  exact  trigonometrical  formulas  just  given  are  tran- 
scendental and  are  therefore  extremely  difficult  to  manipulate.  Great  simplification  is 
obtained  if  the  equations  are  restricted  to  paraxial  rays,  i.e.,  rays  that  make  small  angles 
with  the  optical  axis  and  with  the  normals  to  the  refracting  and  reflecting  surfaces.  The 
paraxial  formulas  given  below  are  deduced  from  the  exact  formulas  by  replacing  the  sines 
of  the  angles  by  the  angles  and  the  cosines  by  unity.  Paraxial  quantities  will  be  indicated 
by  the  corresponding  lower-ease  letter: 

si  -  R  Pi  /i>n 

*i  =  — ^ —  tii  =  -5  MI  <^1) 


-f-  »i  — 


Pi 


R 


The  corresponding  equations  for  a  spherical  mirror  are: 

Pi 


(22) 
(23) 
(24) 
(25) 
(26) 

(27) 

(28) 

(29) 
(30) 

(31) 
(32) 
(33) 

-  +  -  =  -  (34) 

S2         Si          5 

Paraxial  relations  are  linear  and  therefore  are  easily  manipulated  algebraically.    They  are 
very  useful  for  determining  the  approximate  focusing  action  of  an  optical  system. 

If  the  object  is  at  a  very  great  distance,  i.e.,  l/«i  —  0,  the  image  is  located  at  the  second 
focal  point  at  the  distance 

f^-J^—R  (35) 


14-08 


OPTICS 


to  the  right  of  the  apex  A.    If  the  image  is  located  at  a  very  great  distance,  i.e.,  l/s2  =  0, 
then  the  object  is  located  at  the  first  focal  point  at  the  distance 


to  the  left  of  the  apex  A.    For  a  spherical  mirror 

/2  =  /i  =  f 
and  both  focal  points  coincide.    It  should  be  noted  that  in  general 


fi 


(36) 


(37) 


(38) 


This  relation  applies  to  any  optical  system  where  N%  is  the  index  of  refraction  of  the 
image  space  and  NI  that  of  the  object  space. 

MAGNIFICATION.  The  above  relations  suffice  to  locate  an  image  of  an  object,  but 
they  do  not  explicitly  give  any  information  about  the  sizes  of  object  and  image.  As  is 


(a)  MAGNIFICATION    IN   REFRACTIVE  SYSTEMS 


(Jr)  MAGNIFICATION  IN  REFLECTIVE  SYSTEM 
FIG.  7.     Magnification  in  Optical  Systems 

seen  from  the  geometry  of  Figs.  7  (a)  and  7(6),  the  lateral  magnification  which  is  defined 
as  the  ratio  of  image  height  fo>  to  object  height  hi  is 

—  £-g  (39) 

It  is  to  be  noted  that  m  is  negative  for  an  inverted  image  (as  in  Fig.  7)  and  positive  for  an 
erect  image.    Inserting  (39)  into  (11)  there  result  the  very  important  relations 

Ut  =  hiNi  sin  Ui  (40) 


Equation  (40)  or  (41)  is  known  as  Abbe's  sine  condition  and  applies  to  any  number  of  re- 
fracting and  reflecting  coaxial  surfaces  (ATi  index  of  object  space,  N$  index  of  image  space). 
The  corresponding  paraxial  equations 

(42) 


are  known  as  Lagrange's  theorem. 


(43) 


LENSES 
The  longitudinal  magnification  along  the  axis  is 

Wig  ==   "          —  771 

A$i 
The  angular  magnification  is 


u\       m 


The  relation  between  these  magnifications  is 


14-09 

(44) 
(45) 

(46) 


2.  LENSES 

By  applying  the  above  paraxial  relations  to  a  lens  in  air,  Le.,  two  refracting  surfaces,  as 
shown  in  Fig.  8,  it  may  be  shown  that  the  two  focal  points  (Fi,  F2)  and  the  two  principal 
points  (Hi,  #2)  completely  determine  the  paraxial  focusing  characteristics  of  the  lens. 


FIG.  S.     Focusing  Characteristics  of  Thick  Lens  in  Air 

The  first  focal  point,  jPi,  may  be  considered  as  that  object  point  which  is  focused  at  in- 
finity in  the  image  space  so  that  all  paraxial  rays  passing  through  the  point  Fi  are  parallel 
to  the  axis  in  the  image  space.  The  second  focal  point,  P^  may  be  considered  the  image 
of  an  object  at  an  infinite  distance  from  the  lens  in  the  object  space  so  that  all  paraxial 
rays  which  are  parallel  to  the  aris  in  the  object  space  will  pass  through  F*  in  the  image 
space. 

The  distances  indicated  by  /i  and  /i  in  Fig.  8  are  known  as  the  principal  or  equiwlenl 
focal  lengths  and  are  given  by 


J.T  J.fc  1.1*2  » 

/2  "  (N  -  l)[AT(£s  -  Ri)  +  (N  -  l)t]  =  ~h 
The  positions  of  the  focal  points  measured  from  the  refracting  surfaces  are  given  by 


(47) 


(48) 


(49) 


The  focal  distances  &FI  and  s^2  are  known  as  the  front  focal  length  (F.F.L.)  and  back  focal 
length  (B.F.L.)  respectively."  These  focal  lengths  are  easily  measured  by  imaging  an 
object  at  a  great  distance  from  the  lens  and  noting  the  distance  between  image  and  nearest 
surface  of  the  lens  and  then  turning  the  lens  around  and  repeating  the  operation.  The 
positions  of  the  principal  points  Hi  and  Hi  measured  from  the  refracting  surfaces  are 
.given  by 

-MT--/.^^^-  (50) 


AT  -  1    t 


(51) 


14-10  OPTICS 

The  distance  between  the  principal  points  is 


From  (50)  and  (51)  it  is  seen  that 

S^  =  |l  (53) 

SH2         Rz 

In  many  practical  eases  the  lens  thickness,  t,  is  numerically  small  in  comparison  with 
either  Ri,  RZ,  or  (R%  —  -Ki),  and  then 

Rit  Rtf 


For  a  biconvex  lens  with  Rz  =  —  Ri(Ri  positive) 

_        Ri  _    t  _         t 

h  =  2(N  -  1)         SHl  =  2N        *Hz  ~  ~  2N 

If  Rz  7*  RI,  both  principal  planes  move  toward  the  surface  with  the  higher  curvature,  and 
when  RZ  =  ±°°,  i.e.,  for  a  plano-convex  lens, 

The  positions  of  the  focal  and  principal  points  of  an  optical  system  having  been  deter- 
mined, its  focusing  action  may  be  calculated  with  the  aid  of  the  following  formulas  (see 
Fig.  8). 

XiXz  -  fifz  =  -/a2  (56) 

5-5-S  (57; 

mfc__A_:*«_5  (58) 

til  -o-l  72  *i 

sz  =  -Mm  -  1)  (59) 

*i-  ~/2(m^  1}  (60) 

THIN  LENSES.  A  lens  of  negligible  thickness  relative  to  its  focal  length  is  known  as 
a  thin  lens.  For  a  thin  lens,  SHI  —  SH%  —  d  =  t  —  0,  and  great  simplification  in  computa- 
tion results.  For  a  thin  lens  the  distances  si,  sz,  fi,  and/2  in  eqs.  (56)  to  (60)  are  measured 
from  the  (center  of)  lens,  and  the  distance  between  object  and  image  is 


(61) 


COMPOUND  LENSES.  Because  of  the  necessity  for  correcting  aberrations  most 
lenses  are  compound;  i.e.,  they  consist  of  several  lenses  having  a  common  axis.  Some  of 
the  surfaces  of  adjacent  lenses  may  be  in  contact  and  others  are  separated  by  air  spaces. 
The  focal  length  of  two  thin  lenses  in  contact  of  focal  lengths  /aa  and  fa  is 


/.  =  (62) 


The  power  of  a  lens  measured  in  diopters  is  defined  as  P  —  l/f,  where  /  is  measured  in 
meters.    Thus  the  power  of  two  thin  lenses  in  contact  is 

P=*^-  +  ^-=Pa  +  Pb  (63) 

ha         fa 

If  the  two  thin  lenses  are  separated  by  an  air  space  of  thickness  t,  the  resulting  lens  is 
effectively  a  thick  lens  and  the  focal  length  of  the  combination  is 

,  /2o/26  . 

h  =   ,      .    .  -  ;  =  -/i  (64) 

/2o  +  /26  —  t 

and  the  power  is 

P  =   Pa  +  Pb  ~  tPJPl  (65) 


LENSES 


14-11 


/aa  +  /»  - 


The  principal  points  and  focal  points  are  located  at  (see  Fig.  9) 

hat  hbt 

Sffi  = — OET    = 

ha  +  /26   —   t 

SFi  =  /to  +  Afr  -  * 
The  distance  between  the  principal  points  is 

d 


+  /26  - 


(66) 

(67) 


FIG.  9.     Focusing  Characteristics  of  Two  Thin  Lenses  Separated  by  an  Air  Space 

There  are  three  ways  of  compounding  two  lenses. 

1.  Both  lenses  are  converging;  i.e.,  f*a  >  0  and/2&  >  0.    In  this  case  the  focal  length  of 
the  combination  is  a  minirnum  and  the  power  is  a  maximum  when  t  —  0.    The  focal  length 
increases  as  t  is  increased  until  when  t  —  fza  +  f&>  the  focal  length  is  infinite  and  the  system 
is  afocal  or  telescopic.    Objectives,  magnifiers,  and  eyepieces  are  generally  compounded 
of  two  such  lenses  with  t  <  /*a  +  /2&.    The  focal  length  of  the  combination  is  negative  for 
t  >  /2a  +  /25t  and  such  a  combination  forms  the  compound  microscope. 

2.  Both  lenses  are  diverging;  i.e.,  ft*  <  0  and  /2&  <  0.    The  focal  length  of  the  com- 
bination is  negative  and  decreases  in  absolute  values  as  t  is  increased. 

3.  One  lens  is  converging  /Jo  >  0  and  one  diverging  fa  <  0.    If  |  f&  \  <  \  f*a    the  focal 
length  of  the  combination  is  negative  for  values  of  t  <\ha  +f^>\,  it  is  infinite  for  t 
=  1  fza  -h  /2&  [ ;  this  is  the  optical  system  of  the  Galilean  telescope  which  produces  an 
erect  image.    The  focal  length  is  positive  for  t  >  |  /^  -J-  /2j  j  and  corresponds  to  the  opti- 
cal system  of  a  telephoto  lens.    If  j  /#,  |  >  |  /2o  |  the  focal  length  is  always  positive. 

ABERRATIONS.  Actual  imagery  with  practical  lenses  departs  from  the  paraxial 
or  first-order  imagery  discussed  above.  These  departures  are  known  as  aberrations. 
There  are  seven  independent  third-order  aberrations. 

Spherical  aberration  or  aperture  defect  is  illustrated  in  Fig.  10.  In  the  presence  of 
spherical  aberration  the  rays  from  a  point  object  on  the  axis  do  not  reeombine  to  form  a 


CIRCLE  OF 
LEAST  CONFUStON 


OBJECT  POINT 


PARWC1AL 
IMAGE  POINT 


EIG.  10.    Positive  or  Undercorrected  Spherical  Aberration 

point  image  as  required  by  paraxial  theory.  Spherical  aberration  is  generally  measured 
by  the  axial  intersection  distance,  ASz,  which  varies  approximately  as  the  cube  of  the  con- 
vergence angle  Z7i.  Any  simple  convex  lens  has  positive  spherical  aberration  and  is  cor- 
rected by  compounding  it  with  a  concave  lens  which  has  negative  spherical  aberration. 


14-12 


OPTICS 


In  a  single  lens  the  spherical  aberration  is  minimized  if  the  deviation  is  equally  divided 
between  the  two  surfaces  of  the  lens. 

Coma  is  an  extra-axial  aberration;  i.e.,  it  affects  image  points  not  on  the  axis  of  the 
lens.  It  may  be  considered  lateral  spherical  aberration  for  an  extra-axial  point.  If  coma 
is  present  a  point  object  is  imaged  into  a  comet-shaped  figure.  It  increases  directly  with 
the  distance  of  the  object  point  from  the  axis  and  as  the  square  of  the  aperture  of  the  lens. 
Coma,  unlike  spherical  aberration,  may  be  eliminated  from  a  single  thin  lens  for  one  object 
distance;  however,  a  lens  for  zero  coma  is  not  the  same  as  one  for  minimum  spherical 
aberration.  Coma  may  be  eliminated  by  a  stop  of  the  proper  aperture  located  at  the 
proper  distance  from  the  lens.  A  lens  corrected  for  spherical  aberration  and  coma  for  a 
given  object  distance  is  called  an  aplanatic  lens.  If  spherical  aberration  is  absent  the 
condition  for  freedom  of  coma  is  given  by  Abbe's  sine  condition,  eq.  (40)  or  (41). 

Astigmatism,  like  coma,  affects  only  extra-axial  points.  When  astigmatism  is  present 
each  object  point  has  two  images,  one  behind  the  other.  These  images  are  short  lines  at 
right  angles  to  each  other.  One  is  called  the  sagittal  or  radial  astigmatic  line  since  it  is  in 
a  plane  containing  the  axis  of  the  system.  The  other  is  called  the  tangential  or  transverse 
astigmatic  line  since  it  is  at  right  angles  to  a  line  drawn  from  the  lens  axis.  Midway  the 
two  line  images  the  cross-section  of  the  beam  is  a  circle  of  least  confusion.  The  astig- 
matism is  positive  if  the  sagittal  focus  is  farther  from  the  lens  than  the  tangential  focus. 
Otherwise  it  is  negative.  Astigmatism  increases  as  the  square  of  the  distance  of  the  object 
point  from  the  axis  and  directly  as  the  aperture. 

Astigmatism  and  curvature  of  the  field  are  generally  present  together,  and  the  sagittal 
and  tangential  foci  lie  on  curved  surfaces  as  illustrated  in  Fig.  11.  If  only  curvature  of 


L  IN 
1\ 


IDEAL,  IMAGE  PLANE 

S 


(a)  PURE  CURVATURE 
OF  FIELD 


Ct)  POSITIVE    ASTIGMATISM 
WITH  POSITIVE  TANGENTIAL 
AND  SAGITTAL  CURVATURE 


(C)  NEGATIVE  ASTIGMATISM 
WITH  NEGATIVE   TAN- 
GENTIAL AND  POSITIVE 
SAGITTAL  CURVATURE 


FIG.  11.     Astigmatism  and  Curvature  of  Field 

the  field  is  present  the  image  points  lie  on  a  spherical  surface  known  as  the  Petsval  surface, 
which  is  indicated  in  Fig.  11  by  P. 

Distortion  is  the  variation  of  magnification  with  the  distance  of  an  object  point  from 
the  axis.  Positive,  or  "barrel"  distortion,  exists  if  the  magnification  decreases  with  in- 
creasing distance  of  an  object  point  from  the  axis.  Negative,  or  "pincushion,"  distortion 
exists  if  the  magnification  increases  with  increasing  distance  of  an  object  point  from  the 
axis.  Distortion  increases  as  the  cube  of  the  distance  of  an  object  point  from  the  axis. 

Chromatic  aberration  is  due  to  the  fact  that  the  focal  length  of  a  lens  depends  upon 
its  index  of  refraction  and  thus  upon  the  wavelength  of  the  light  used.  As  a  consequence 


FIG.  12.     Longitudinal  and  Lateral  Chromatic  Aberration 

a  single  lens  forms  a  series  of  colored  images  of  different  magnifications  at  different  dis- 
tances from  the  lens  (Fig.  12) .  The  variation  of  image  distance  with  wavelength  is  known 
as  kmg&todinal  chromatic  aberration.  The  difference  Sac  —  S&  is  taken  as  a  measure  of 


LENSES 


14-13 


the  longitudinal  aberration,  and  for  a  thin  lens  it  is  given  by 

1 


S2C   —    -S2F   : 


(-  +  — 
\AD 


(69) 


where  V  is  given  by  eq.  (6).     The  variation  of  image  size  with  wavelength  of  refraction  is 
known  as  lateral  chromatic  aberration  and  is  measured  by  he  —  ^F- 

STOPS.  Any  obstacle  which  limits  the  rays  that  can  be  transmitted  through  an 
optical  system  is  a  stop.  Stops  serve  (1)  to  limit  the  aperture  of  any  bundle  of  rays,  and 
thus  the  amount  of  light,  that  reaches  any  image  point,  and  (2)  to  limit  the  extent  of  the 
object  which  is  imaged,  or  the  field  of  view.  The  stop  that  limits  the  diameter  of  the  bundle 
of  rays  that  can  enter  the  optical  system  is  called  the  aperture  stop  or  iris.  The  stop  that 
limits  the  field  of  view  is  known  as  the  field  stop.  In  Fig.  13,  NP  is  the  image  of  AS  formed 


OBJECT  PLANE 


IMAGE  PLANE 


CIRCLES  OF  CONFUSION 
OF  RADIUS -a 


FIG.  13.     Aperture  Stop,  AS,  and  Field  Stop,  FSi,  in  an  Optical  System 

by  that  part  of  the  optical  system  Gens  I/i)  situated  between  AS  and  the  object.  Since 
NP  limits  the  angular  aperture  Ui  of  the  beam,  AS  is  the  aperture  stop.  NP  is  known  as 
the  entrance  pupil.  If  an  actual  stop  were  located  at  NP  instead  of  AS,  then  the  iris  and 
entrance  pupil  would  coincide.  The  image  XP  of  AS  formed  by  that  part  of  the  optical 
system  (lens  L*)  situated  between  AS  and  the  image  is  known  as  the  exit  pupil.  The 
angle  subtended  by  the  object  at  the  center  of  the  entrance  pupil  is  known  as  the  angular 
field  of  view  of  the  object.  In 
Fig.  13  the  angular  field  of  view 
of  the  object  is  limited  to  21^1 
by  the  aperture  stop  FSi.  If 
it  is  desirable  to  have  a  sharp 
boundary  of  the  field  of  view 
it  is  necessary  to  have  FSi  in 
the  plane  of  the  object. 

Stops  play  a  very  important 
role  in  eliminating  those  rays 
that  would  produce  excessive 
aberrations. 

DEPTH  OF  FOCUS.  The 
maximum  total  distance,  A&2, 
that  a  viewing  screen  may  be 
moved  along  the  axis  of  an 
optical  system  to  produce  a 
circle  of  confusion  of  prescribed 
radius,  r,  is  called  the  depth  of 
focus  (Fig.  14a) .  It  is  given  by 


4x)  DEPTH  OF  FOCUS 


CIRCLE  OF  CONFUSION 
OF  RADIUS yv 


t=*2r—  =  2rcot  C72     (70) 


(ti  DEPTH  OF  FIELD 
FIG.  14.    Difference  between  Depth  of  Field  and  Depth,  of  Focus 


where  Sz  is  the  image  distance, 
at  best  focus,  measured  from 
the  exit  pupil  of  radius  a,$. 

DEPTH  OF  FIELD.  The  maximum  total  distance,  ASi,  that  an  object  may  be  moved 
along  the  axis  to  produce,  at  a  fixed  image  distance,  a  circle  of  confusion  of  prescribed 
radius,  r,  is  called  the  depth  of  field  (Fig.  146).  It  is  given  by 

_•* 


-f- 


(71) 


14-14 


OPTICS 


where  m  is  the  magnification  and  Si  is  the  object  distance,  at  best  focus,  measured  from 
the  entrance  pupil  of  radius  a\.    If  r  <$C  mai,  then 


ASi  «  2  -  — 


2  -  cot 


(72) 


3.  PHOTOMETRY 

The  problem  in  photometry  is  to  obtain  a  quantitative  evaluation  of  radiant  flux  with 
respect  to  its  capacity  to  produce  the  sensation  of  brightness.  The  sensation  of  brightness 
evoked  by  a  given  amount  of  radiant  energy  is  different  for  different  individuals  and  is 
different  for  the  same  individual  under  different  conditions  of  observation.  Equal  amounts 
of  radiant  energy  per  unit  wavelength  interval  throughout  the  visible  spectrum  do  not 
produce  visual  sensations  of  equal  brightness.  A  luminosity  curve  shows  as  ordinates 
the  relative  effectiveness  of  various  wavelengths  to  evoke,  for  a  particular  observer, 
visual  sensations  of  equal  brightness.  Figure  15  shows  two  luminosity  curves  applying  to 


t.O 


?   0.8 


0.7 


0.6 


> 
<   0.5 


0.4 


C  03 

02 
0.1 


I 


\ 


\ 


\ 


A 


\ 


380  400  420  440  440  48O  500  520  540  560  580  600  620  640  660  680  700  720  740 
X  IK  MILLIMICRONS 

FIG.  15.    Curve  (a),  Standard  Luminosity  Function,  Applying  to  Normal  Vision  with  Good  Lighting 
Conditions;  Curve  (&),  Luminosity  Function  Applying  to  Vision  at  Very  Low  Light  Levels 

photopic  or  normal  vision  (at  high  light  levels)  and  to  scotopic  vision  (at  very  low  light 
levels).  These  are  average  luminosity  curves  obtained  for  a  number  of  observers.  The 
use  of  many  luminosity  curves  would,  obviously,  lead  to  endless  confusion  and  ambiguity 
in  photometric  measurements  and  specification.  To  avoid  this,  the  International  Com- 
mission on  Illumination  adopted  Fig.  15a  as  the  standard  luminosity  curve.  It  is  important 
to  realize  that  the  standard  luminosity  curve  is  essentially  an  arbitrarily  assumed  standard 
for  purposes  of  standardization  and  specification  of  photometric  data  and  is  not  neces- 
sarily the  retinal  response  of  any  individual.  Table  3  gives  the  standard  luminosity  func- 
tion, y(\)  of  Fig.  I5at  at  10-millimicron  intervals. 

Table  3.    Standard  Luminosity  Function  1 


X  in  m/i 

Z/(X) 

X  in  m/t 

PCX) 

Xin  mjt 

PCX) 

X  in  mjt 

»(X) 

380 

0.0000 

480 

0.139 

580 

0.870 

680 

0.0170 

390 

.0001 

490 

.208 

590 

.757 

690 

.0082 

400 

.0004 

500 

.323 

600 

.631 

700 

.0041 

410 

.0012 

510 

.503 

610 

.503 

710 

.0021 

420 

.0040 

520 

.710 

620 

.381 

720 

.0010 

430 

.0116 

530 

.862 

630 

.265 

730  • 

.0005 

440 

.0230 

540 

.954 

640 

.175 

740 

.0003 

450 

.0380 

550 

.995 

650 

.107 

750 

.0001 

460 

.0600 

560 

.995 

660 

.061 

760 

.0001 

470 

.0910 

570 

.952 

670 

.032 

PHOTOMETRY 


14-15 


Table  4  gives  the  names,  symbols,  and  basic  mks  units  of  radiometry  and  photometry 
as  recommended  by  the  committee  on  colorimetry  of  the  Optical  Society  of  America. 
The  term  "luminance"  hi  Table  5  replaces  the  older  term  "brightness"  which  led  to  con- 
fusion between  the  objective  concept  of  brightness  as  a  measurable  quantity  and  the  sub- 
jective concept  of  brightness  which  refers  to  the  sensation  in  the  consciousness  of  the 
human  observer.  It  is  recommended  that  the  term  "brightness"  be  used  only  in  the  latter 
sense. 

*  Table  4 


Radiometry 

Photometry 

Name 

Sym- 
bol 

Unit  MKS 

Name 

Sym- 
bol 

Unit  MKS 

]R^.Hi?mt  energy 

U 

P 
W 

J 

N 
H 

Joule 
Watt 
Watt/m2 
Watt'w  (steradian) 
Watt  "wxm2 
Watt/m2 

Luminous  energy  

TAirninous  flux 

Q 

F 

\ 

B 
E 

Talbot 
Lumen 
Lumen  ms 
Lumen  e*  (candle) 
Lumen  Wxm2 
(candle  m*) 
Lumen  7m2  flux) 

Radiant  flux  

Radiant  emittance.  .  . 
Radiant  intensity  
Radiance  

Luminous  emittanee.  . 
Luminous  intensity  .  .  . 
Luminance 

Irradiance  

TH«minfl.nee  

The  ratio  of  any  photometric  quantity  to  the  corresponding  radiometric  quantity  in 
Table  5  is  equal  to  the  absolute  luminosity  or  luminous  efficiency  (generally  expressed  as 
lumens  per  watt)  of  the  radiant  energy.  Thus  1  watt  of  monochromatic  radiant  flux  of 
wavelength  555  mju  (corresponding  to  the  peak  of  the  standard  luminosity  curve)  is 
equivalent  to  685  lumens.  This  efficiency  of  685  lumens  per  watt  is  based  on  the  fact  that 
the  new  proposed  international  photometric  standard  of  a  black  body  at  the  temperature 
(2043.8  deg  K)  of  freezing  platinum  shall  have  a  luminance  of  6  X  10*  candies  per  m* 
(60  candles  per  cm2). 

Table  5.    Conversion  Factors  for  Units  of  Dltuninance 


Multiply' 
Number 


to 
Obtain 


Lux 


Foot-candle 


Phot: 


Mffliphot 


Lumen/m2      Lux , 

Lumen/ft2       Foot-candle. 

Lumen/cm2    Phot 

MilHphot. . . 


0.0929 
0.0001 
.  1 


10.76 


0.001076 
1.076 


10,000 

929 

I 

1,000 


10 

0.929 
0.001 
1 


If  the  radiant  flux  is  monochromatic  its  luminous  efficiency  is  simply  6&5£(X).  If  the 
radiant  flux  is  not  monochromatic  but  consists  of  a  continuous  spectrum,  then  if  P(X)  is  the 
radiant  flux  per  unit  wavelength  (watts  per  millimicron)  the  luminous  efficiency  is 


dX 


685- 


(73) 


Luminous  efficiency  should  not  be  confused  with  the  efficiency  of  a  practical  light 
source,  which  is  the  ratio  of  the  total  luminous  flux  to  the  total  power  input.  The  effi- 
ciency of  a  source  of  light  is  less  than  K  since  generally  a  fraction  of  the  total  power  input 
is  not  converted  into  radiant  flux. 

LUMINOUS  INTENSITY  OR  CANBLEPOWER  OF  A  SOURCE.  By  a  source  win 
be  understood  (a)  any  self-luminous  object,  such  as  an  incandescent  body,  or  (6)  any 
illuminated  object  which  so  completely  diffuses  (either  by  reflection  or  transmission)  the 
incident  light  that  it  acts  as  a  source. 

The  intensity  of  a  source  is  defined  as 

I  =  *?  (74) 


and  is  measured  in  lumens  per  steradian,  or  candles.    In  Fig.  16,  let  the  small  plane  source 


14-16 


OPTICS 


of  area  A0  emit  P  watts  or  F  -  685  /*°°£(X)P(X)  d\  lumens.    The  two  small  -receivers  of 

•'O 

areas  Al  and  A2  subtend  the  same  solid  angle  Aco  at  the  (say,  center  of)  source  and  are 
located  at  the  same  distance  D  from  the  source.  The  number  of  lumens  AFi  and  AF2 
contained  in  the  same  solid  angle  Aco  will,  in  general  be  different.  Thus  the  intensity 

— — -  ,  is  not  equal  to  the  intensity,  /«  =  -r—  ,  measured  along 

AOJ  ™  **• 


normal  to  the  source,  70 


AF2 


Aco  ^ 

a  direction  at  the  angle  a. 
with  the  normal.  In  specify- 
ing the  intensity  or  candle- 
power  of  a  source,  it  is  there- 
fore necessary  to  state  the 
direction  in  which  the  inten- 
sity was  measured  or  give 
a  candlepower  distribution 
curve.  The  directional  char- 
acteristic of  many  extended 
sources  follow  Lambert's  law, 
which  states  that 

Ia  s*  Jo  cos  a.      (75) 

With    a   uniform    point    or 
spherical  source,  the  inten- 
sity is  independent  of  direc- 
tion and  is  F/4ir.    Thus  a 
uniform  point  or  spherical 
source  of  an  intensity  of  1 
candle  emits  4?r  lumens. 
ILLUMINANCE.     A  surface  of  area  dA,  placed  in  a  field  of  flux,  is  illuminated  with 
the  illuminance 

jn 

E  —  -—  (lumens  per  m2  in  mks  units)  (76) 

dA 

where  dF  is  the  flux  incident  on  the  surface.    The  illuminance  of  receiver  A\  (Fig.  16)  is 

AP,       JoAco        Jo  (Ai  cos  0)        IQ 

(77  a) 


FIG.  16.     Diagram  Illustrating  Some  Photometric  Concepts 


The  illuminance  of  receiver  A  2  is 


,  =  —z  cos  0  —  —  cos  a.  cos  0 


In  normal  incidence 


(776) 
(78) 


which  is  the  well-known  inverse-square  law.  The  inverse-square  law,  which  is  the  basis 
of  most  visual  photometers,  applies  to  large  extended  sources,  if  D  is  considerably  larger 
than  the  extension  of  the  source.  Thus  (78)  holds  to  within  about  1  per  cent  if  D  is  at 
least  5  times  the  greatest  linear  dimension  of  the  source. 

LUMINANCE.  The  luminance,  B,  of  the  surface  AQ  (Fig.  16)  in  any  direction  is  the 
ratio  of  the  intensity  Ia  in  that  direction  to  the  area  of  the  projection  of  AQ  on  a  plane 
perpendicular  to  the  direction,  i.e., 

(79) 


^-0  COS  Oi 

Luminance  is  measured  in  candles  per  square  meter  in  mks  units.  If  Lambert's  law  is 
followed,  then 

s  =  4°J21£  .  Jj  _  Bo  (so) 

AQ  COS  CX.          AQ 

and  the  luminance  of  a  surface  is  independent  of  a.  The  brightness  sensation  when  ob- 
serving a  surface,  whether  self-luminous  (as  the  luminescent  screen  of  a  cathode-ray  tube) 
or  diffusely  transmitting  or  reflecting  (as  a  television  or  movie  projection  screen),  depends 
upon  the  luminance  of  the  surface.  Hence  if  the  surface  obeys  Lambert's  law,  its  lumi- 
nance is  the  same  in  all  directions,  and  it  will  appear  equally  bright  from  whatever  angle 
it  is  viewed. 


LIGHT  MEASUREMENT 


14-17 


LUMINOUS  EMITTANCE.    The  luminous  emittanee,  L,  of  a  surface  is  the  total 
luminous  flux  emitted  per  unit  of  area,  or 

i  -  |  (8D 

Luminous  emittanee  is  measure  in  lumens  per  square  meter  in  mks  units. 
The  luminous  emittanee  of  a  surface  obeying  Lambert's  Jair  is  found  to  be 

L  =  icB  (82) 

Hence  a  perfectly  diffusing  (emitting,  transmitting,  or  reflecting)  surface,  whose  luminance 
is  B  candles  per  square  meter  has  a  luminous  emittanee  of  xB  lumens  per  square  meter. 
Or  a  perfectly  diffusing  surface  of  luminance  B  and  area  A  emits 


(83) 


F  «  TrBA  lumens 
If  the  surface  radiates  on  both  sides  with  luminance  B,  then 

F  =  2TBA  (84) 

PHOTOMETRIC  UNITS.  Equation  (82)  is  the  basis  for  another  unit  of  luminance 
called  meter-lambert.  By  definition  1  meter-Lambert  is  the  luminance  of  a  perfectly  dif- 
fusing surface  emitting,  reflecting,  or  transmitting,  1  lumen  per  m2.  Thus  1  meter4ambert 
equals  1/x  candle  /m2.  This  unit  (meter-lambert)  is  very  convenient  when  dealing  with 
non-self-luminous  surfaces  such  as  perfectly  diffusing  (transmitting  or  reflecting)  surfaces. 
If  these  surfaces  do  not  absorb  any  light,  the  number  of  lumens  incident  on  them  is  equal 
to  the  number  of  lumens  transmitted  or  reflected.  Hence  the  illuminance  in  lumens  per 
square  meter,  the  luminous  emittanee  ia  kirnens  per  square  meter,  and  the  luminance  in 
meter-lamberts  are  all  numerically  equal.  If  these  surfaces  do  absorb  some  light,  the 
luminance  in  meter-lamberts  is  equal  to  the  iUuminance  in  lumens  per  square  meter 
multiplied  by  the  fraction  of  the  incident  light  that  is  transmitted  or  reflected.  The 
luminous  emittanee  in  lumens  per  square  meter  is  still  equal  to  the  luminance  in  meter- 
lamberts.  If  the  surface  does  not  obey  Lambert's  law  its  luminance  depends  upon  the 
angle  of  observation  and  the  advantage  of  this  unit  disappears.  The  luminance  of  such 
a  surface  in  a  particular  direction  in  meter-lamberts  may  then  be  interpreted  as  the  mim- 
ber  of  lumens  per  square  meter  that  a  perfectly  diffusing  surface  of  the  same  luminance 
would  radiate. 

Besides  the  mks  units  given  above,  there  are  many  others  in  widespread  use.  They 
differ  from  the  mks  units  only  in  being  based  on  different  units  of  area.  Tables  5  and  6 
give  the  names  and  the  relative  magnitudes  of  the  more  common  units  of  illuminance  and 
luminance. 

Table  6.    Conversion  Factors  for  Units  of  Luminance 


Multiply 

^N,    JS^ 

Foot- 

>v     ^S. 

Candfe 

Candle 

Candle 

Candle 

Miffi- 

lambert 

Meter- 

X>Os. 

per 

per 

per  ft2 

per 

Lambert 

kmbert 

(equivaieat 

binbert 

cm 

m.2 

m2 

fatir 

Obtain     ^o^ 

4-            * 

candk) 

Candle  per  cm2  (Stilb) 

1 

0.1550  ; 

0.0010764 

!Q-« 

0.3183 

0.0003183 

0.0003426 

0.00003183 

Candle  per  in.2 

6.452 

I 

0.006944 

6.452 

2.054 

0.002054 

0.00221 

6.0032054 

xw-* 

Candle  per  ft2 

929 

144 

I 

0.0929 

295.7 

0.29^ 

0.3183 

0.02957 

Candle  per  m2     .... 

10,000 

1,550 

^10.764 

1 

3,183 

3.183 

3.426 

0.3183 

Lambert  (cm4ambert) 

3.142 

0.4869 

3.382 

3.142 

1 

0.001 

0.001076 

XIQ-* 

xw~4 

MilHlambert 

3,142 

486.9 

3.382 

0.3142 

1,000 

I 

1.0764 

O.I 

Foot-lamberfc  

2,919 

452.4 

3.142 

0.2919 

929 

0.929 

0.0929 

Meter-lambert  

31,420 

4,869 

33.82 

3.142 

10* 

10 

10.76 

I 

4.  LIGHT  MEASUREMENT 

The  methods  of  light  measurement  may  be  divided  into  two  classes;  visual  photometry 
and  physical  photometry. 

In  visual  photometry  the  human  eye  is  the  detector.  Although  the  human  eye  is 
incapable  of  measuring,  it  is  capable  of  fairly  accurately  judging  the  equality  of  lumi- 
nances of  adjacent  areas.  In  a  visual  photometer,  two  adjacent  areas  of  a  screen  are 


14-18 


OPTICS 


illuminated  by  a  calibrated  source  and  an  unknown  source.  The  observer  adjusts  the 
illuminance  on  the  half-field  produced  by  the  calibrated  source  (by  varying  the  distance 
between  source  and  screen)  until  he  judges  the  two  half-fields  to  be  equally  "bright." 
The  better  visual  photometers  (such  as  the  "Macbeth  Illuminometer")  use  a  Lummer 
Brodhun  cube  to  split  the  field.  Relatively  accurate  measurements  may  be  made  with 
visual  photometers  only  if  the  spectral  distribution  of  the  calibrated  and  unknown  sources 
are  approximately  the  same.  Although  a  series  of  niters  may  relieve  this  situation,  meas- 
urements upon  sources  of  different  colors  (heterchromatic  photometry)  are  generally 
subject  to  great  errors  unless  a  flicker  photometer  is  used. 

In  physical  photometry,  the  detector  is  generally  a  photovoltaic  or  photoemissive  cell. 
A  cell  will  give  true  photometric  values  regardless  of  the  color  of  the  light  if  corrected 
with  a  suitable  filter,  so  that  its  sensitivity  throughout  the  spectrum  is  proportional  to 
the  luminosity  curve  of  Fig.  15a.  The  most  important  error  in  physical  photometry  is 
generally  due  to  the  difference  between  the  spectral  response  of  the  cell  and  the  standard 
luminosity  curve.  Other  errors  arise  from  the  directional  and  temperature  characteristics 
of  cells.  Most  commercial  light  measurements  are  now  made  using  a  photovoltaic  cell 
with  an  "eye"-corrected  filter. 


5.  PHOTOMETRIC  RELATIONS  IN  NON-VISUAL  OPTICAL 

SYSTEMS 

One  of  the  important  performance  characteristics  of  an  optical  system  is  its  light- 
gathering  power.    In  Fig.  17  Ihe  small  object  of  area  AI  radiates  in  all  directions,  but  the 


FIG.  17.    Diagram  Illustrating  Some  Photometric  Relations  in  Optical  Systems 

lens  accepts  and  focuses  only  a  fraction  of  the  flux  on  the  image  of  Area  A%.  If  A\  emits 
according  to  Lambert's  law,  then  the  fraction  of  the  flux  that  the  lens  accepts  is 

e  =  sin2  Ui  (85) 

where  U\  is  the  angle  subtended  by  the  radius  of  the  entrance  pupil  at  A\.  The  quantity  e 
might  be  called  the  (geometric)  efficiency  of  the  optical  system.  If  B i  is  the  luminance  of 
A\j  then  the  total  flux,  F,  that  A\  emits  is  irBiA\  and  the  flux  accepted  by  the  lens  is 

Fi  =  vBiAi  sin2  Ui  (86) 

If  the  lens  has  no  losses,  FI  is  the  flux  that  will  reach  the  image.  If  there  are  losses  in  the 
optical  system,  let  k  be  the  fraction  of  the  incident  light  that  it  transmits,  and 

Fz  =  kFi  =  irBiAik  sin2  Ui  (87) 

The  quantity  fc  sin2  Ui  might  be  called  the  effective  efficiency  of  the  optical  system. 
The  illuminance  of  the  image  is 


ik  sin2 


A* 


m? 


ik  sin2  Uz 


(88) 


This  equation  is  the  basic  relation  giving  the  flux  density  on  a  small  area  (.4.2  in.  Fig.  7)  of 
the  image,  located  on  the  axis  of  an  aplanatic  optical  system,  produced  by  an  extended 
source  emitting  according  to  Lambert's  law. 


PHOTOMETRIC  RELATIONS 


14-19 


.8 

7 

\ 

.5 

\ 

I 
1-       .2 

Z 

U-               *Q 

\ 

> 

\ 

_.     .07 

o   -06 

u-    .05 

z  *04 

3    .03 
u. 
0    .OZ 

o 

Z 

£I007 
tt.006 
^  .OO5 
«>.004 

•OOS 
.002 

\ 

\ 

\ 

\ 

\ 

\ 

L 

\ 

\ 

\ 

\ 

a              .2      .3    .4  -S  .  6,7.83  UO            2        3     4-  5  678<UO           » 

FIG.  18.     Efficiency  of  a  Lens  Used  for  Projection  at  Infinite  Tnrow  as  a  Function  of  F/Number 


Most  lenses  are  rated  (as  to  their  light-gathering  power)  by 

1 


and  sometimes  by 


//number  =/n-        = 


Relative  aperture  —  R  A  —  — 


(89) 
(90) 


Microscope  objectives  are  rated  by 

Numerical  aperture  =  NA  —  Nj.  sin  Ui  (91) 

If  the  entrance  and  exit  pupils  coincide  with  the  principal  planes  of  the  optical  system, 
eq.  (88)  may  be  written 

}&       S-j?       irBi  -  \RA}  /oo\ 

•nt     i>  j» i_    ==   I- \*>-£) 

j£z  -  fjstf  ^  +  ^  ^         4    K  (m2/4)  (22^)2  +  (m  -  l)s 

For  camera  and  telescope  objectives,  the  object  distance,  Si,  is  usually  very  large  and  the 
image  distance,  Sj,  is  approximately  equal  to  the  focal  length  (or  m  ^  0)  so 

«£?!  JL  (93) 


If  the  relative  aperture  is  small  or //number  large,  then  for  any  magnification 


For  high-efficiency  projection  systems,  it  is  preferable  to  define //number  as 

1 


(95) 


14-20 


OPTICS 


instead  of  eq.  (89).  For  values  of  U/  less  than  about  10°,  sin  C7>  -  tan  Uf  and  the  two 
definitions  agree,  but  they  disagree  for  larger  values  of  £//.  The  efficiency  of  a  projection 
lens  for  infinite  throw  is  thus 

0  25 
eM  =  sin2  Uf  =  T^T.  (96) 


For  a  lens  immersed  in  air,  the  smallest  FN  possible  is  0.5,  since  then  the  efficiency  is 
unity,  and  all  the  light  emitted  is  accepted  by  the  lens.    For  a  lens  immersed  in  a  medium 

of  index  NI  the  efficiency  is  e^  =  NJ  sin2  Uf  and  the  FN  is  ;— — •      •  • .    Figure  18  shows 

the  efficiency  ex  of  a  lens  as  a  function  of  //number.    It  is  seen  that  the  efficiency  of  most 
lenses  is  very  low. 

The  efficiency  of  a  given  optical  system  decreases  when  the  magnification  or  throw 
decreases.    The  efficiency  at  a  magnification  ra  is 

7)2  * 

. °° 


1  +  (1  - 


c1 


-  2m 


Thus  an  ordinary  lens  with  an  FN  of  2  has  an  ««,  of  6.25  per  cent  and  an  em  of  about  4.6 
per  cent  when  used  at  a  magnification  of  —  6.  Similarly  a  lens  with  an  e^  of  25  per  cent 
(FN  =  1)  will  have  an  efficiency  of  19.6  per  cent  at  a  magnification  of  —  6. 

The  illuminance  of  an  extra-axial  area  (Awt  Fig.  17)  produced  with  a  lens  of  small 
aperture  is  approximately  given  by 

*•- 2*  «***£=&  (98) 

•where  w  is  the  field  angle  (see  Fig.  17) . 


6.  REFLECTIVE  OPTICAL  SYSTEM  FOR  TELEVISION  PROJECTION 

Two  important  requirements  of  an  optical  system  for  television  projection  are:  (1)  that 
it  be  capable  of  focusing  a  large  field  (large  tube  face)  and  (2)  that  it  have  high  efficiency. 
The  need  for  a  large  field  is  that,  owing  to  current  saturation  and  heating  of  the  luminescent 
screen,  the  light  output  from  a  cathode-ray  tube  increases  with  the  size  of  the  tube,  for  a 
given  beam  power  input. 

The  most  efficient  optical  systems  capable  of  focusing  considerable  fields  are,  in  general, 
of  the  reflective  type.  One  of  the  simplest  (and  best)  of  these  consists  of  a  spherical  mirror 
and  an  aspherical  aberration-correcting  lens  located  at  the  center  of  curvature  of  the 
mirror  (Fig.  19). 

.FRONT  FACE 
SPHERICAL 
MIRROR 


FIG.  19.     Reflective  Optical  System  for  Television  Projection 


The  aberrations  of  any  optical  system  with  a  high  geometrical  efficiency  can  be  suffi- 
ciently corrected  for  only  one  position  of  object  and  image.  As  a  result  a  given  system  is 
good  for  only  a  small  range  of  magnification,  and  different  systems  have  to  be  designed  for 


REFLECTIVE  OPTICAL  SYSTEM 


14-21 


different  magnifications  or  throws.    A  highly  efficient  system  has,  because  of  the  large 
convergence  angles,  relatively  shallow  depth  of  focus  and  depth  of  field;  see  eos.  (70)  and 

FOCUSING  WITH  MIRROR  ANB  CORRECTING  LENS.  Because  of  the  symmetry 
of  tne  spnere,  an  optical  system  consisting  of  a  spherical  mirror  and  a  small  aperture  looted 
at  the  center  of  curvature  of  the  sphere  suffers  from  only  two  aberrations:  spherical  aber- 
ration which  is  uniform  all  over  the  field,  and  curvature  of  the  field.  With  large  apertures, 
extra-axial  aberrations  of  higher  order  enter  the  field,  but  they  are  not  too  large  (see 
Fig.  20). 

The  purpose  of  the  correcting  lens  is  to  correct  for  the  spherical  aberration  of  the  mirror 
without  introducing  any  serious  aberrations  of  itself.  This  is  accomplished  by  making 


FIG.  20.     Image  Properties  of  System  Consisting  of  Spherical  Mirror  and  Aperture  Located  at  Center 

of  Curvature 

the  lens  as  weak  as  possible  and  locating  it  in  the  plane  of  the  aperture  at  the  center  of 
curvature.    In  this  way  the  symmetry  property  of  the  spherical  mirror  is  least  disturbed. 
The  curvature  of  the  field  is  not  corrected  as  it  is  actually  used  to  good  advantage  in 
cathode-ray-tube  projection. 
Equation  (20)  may  be  written 

JL+  I=_ 

Pi       PS  xt  JUQ 

where  /Z70  is  the  focal  length  of  a  zone  of  the  mirror  of  aperture  sin  U&.  The  zonal  focal 
length  fu0  thus  increases  with  the  aperture  of  the  zone. 

The  spherical  aberration  of  the  mirror  may  be  interpreted  as  focusing  by  means  of 
zones,  each  zone  having  a  different  focal  length.  The  correcting  lens  has  to  be  such  that 
each  zone  of  the  lens  has  a  different  focal  length,  compensating  for  the  various  focal  lengths 
of  the  mirror  and  resulting  in  a  focusing  system  with  all  zones  of  the  same  focal  length. 

The  shape  of  the  correcting  lens  will  thus  depend  upon  the  zonal  focal  length  of  the 
mirror  one  chooses  as  the  focal  length  of  the  optical  system  (mirror  plus  correcting  lens). 
Since  theoretically  there  are  an  infinite  number  of  zones  on  the  mirror,  there  are  theo- 
retically an  infinite  number  of  correcting  lens  shapes  that  will  produce  a  system  in  which 
all  zones  have  the  same  focal  length, 

Since  the  mirror  with  an  aperture  at  the  center  of  curvature  has  no  extra-axial  or  chro- 
matic aberrations,  such  aberrations  are  caused  by  the  correcting  lens  itself,  i.e.,  by  the 
power  or  slopes  on  the  correcting  lens.  From  the  standpoint  of  these  aberrations,  there- 
fore, that  shape  should  be  chosen  whose  maximum  slope  is  the  least.  Thus  if  the  paraxial 
(central)  focal  length  of  the  mirror  is  chosen  as  that  of  the  system,  then  the  central  focal 
length  of  the  correcting  lens  is  infinite  and  the  shape  of  the  curve  is  concave.  If  a  zonal 
focal  length  of  the  mirror  is  chosen  as  that  of  the  system  there  will  be  a  zonal  focal  length 
of  the  correcting  lens  which  is  infinite  and  the  shape  of  the  curve  is  convex  at  the  ce&ter 


14-22 


OPTICS 


and  concave  past  this  zone.    If  a  peripheral  focal  length  is  chosen,  the  required  correcting 
lens  is  convex.    The  maximum  slope  is  least  for  a  convex-flat-concave  curve. 

The  shape  of  the  correcting  lens  must  be  such  that  all  rays  emanating  from  an  object 
point  0i,  and  reflected  by  the  mirror,  shall  meet  at  the  image  point  Oi  located  at  a  dis- 
tance S  from  the  correcting  lens.  Figure  21  shows  three  rays  emanating  from  Oi  and 


REFERENCE 
ZOME 


-MIRROR 


SINE  U0 


FIG.  21.    Diagram  Illustrating  the  Effect  of  the  Correcting  Lens 

striking  the  mirror  at  different  apertures.  Without  the  presence  of  the  correcting  lens, 
rays  1,  2,  3  would  intersect  the  axis  at  distances  Pz,  Pz,  and  P^  from  the  center  of  curvature. 
The  slopes  on  the  correcting  lens  have  to  be  such  (approximately  as  shown  on  Fig.  21) 
that  all  three  rays  intersect  at  Oa;  hence,  the  correcting  lens  has  a  flat  zone  at  the  point 
where  ray  2'  passes,  negative  slope  where  ray  1'  passes,  and  positive  slope  where  ray  3' 
passes. 

From  the  point  of  view  of  spherical  aberration,  if  the  zone  where  ray  2  strikes  the  mirror 
is  taken  as  a  reference,  then  the  mirror  has  negative  spherical  aberration  for  smaller  aper- 
tures and  thus  requires  a  positive  lens  for  correction,  and  positive  spherical  aberration  for 
larger  apertures  and  thus  requires  a  negative  lens. 


ZONE   HAVING 
FOCAL  LENGTH 
OF  SYSTEM 


CORRECTOR    FOR   HIGH    MAGNIFICATION 


FIG.  22.    Diagram  Showing  Why  the  Aperture  of  the  Correcting  Lens  Depends  upon  Magnification 

The  shape  and  size  of  the  correcting  lens  depend  upon  the  throw  or  magnification  for 
which  the  system  is  to  be  used.  For  a  given  focal  length  and  relative  aperture  the  cor- 
recting lens  aperture  decreases  as  the  magnification  decreases  (see  Fig.  22).  That  this 
must  be  so  may,  be  surmised  from  the  fact  that  for  unity  magnification  the  lens  aperture 
object  and  image  coincide  at  the  center  of  curvature.  Figure  23  shows  the 


REFLECTIVE  OPTICAL  SYSTEM 


14-23 


variation  of  correcting  lens  semiaperture  and  mirror  semiaperture,  with  magnification, 
for  a  system  of  given  focal  length  and  efficiency.  All  distances  in  Fig.  23  are  measured  in 
terms  of  the  radius  of  curvature  of  the  mirror;  i.e.,  the  radius  of  curvature  is  taken  as  the 
unit  of  length. 

The  focal  length  of  the  complete  optical  system  depends  upon  the  shape  of  the  correcting 
lens.  In  general,  the  higher  the  efficiency  for  which  a  system  is  designed  the  greater  the 
focal  length  (in  terms  of  R).  Thus  for  an  extremely  inefficient  system  (requiring  only  a 
small  aperture  at  the  center  of  curvature  and  no  correcting  lens)  the  focal  length  would 

<£~-.456 


< 

I 
u 
w 


\ 

1 

Acm 
km 

1   111               i     i 

=SEM1  APERTURE  OF  CORRECTING 

X        «                        »                      W                       B 

-     *              "             H    MIRROR  FOR 

LENS        [ 
•         MINIMU1 
AXIAL  POf  NT 
ttJSHWUM  RAY  =.3. 

u 
317= 

^ 

\ 

=  SI 

N  Uc 

!   f  I 

X 

X 

x 

V 

**- 

•  „, 

.*• 

•  —  j 

.1  

—  — 
^.  — 

..    • 

—  - 

_- 

•» 

»  .  • 



_ 

L 

y, 

' 

' 

,' 

""" 

/ 

/ 

/ 

I/ 

\                  Z          3      4    567*9  tO                2         3      4     56789  100              Z          3     4     i    «7  491000 

m 

FIG.  23.     Variation  of  Semi-aperture  of  Mirror  and  Correcting  Lens  with  Magnification 

be  0.522,  whereas  a  reasonably  good  design  of  a  system  with  an  efficiency  of  about  50  per 
cent  (requiring  a  correcting  lens)  will  have  a  focal  length  of  about  0.53B. 

Relations  between  the  throw  5,  magnification  m,  object  distance  Pjy  and  the  focal  length 

/aie:  fl-**  (100) 

PI  =  /(m  +  1)  (ioi) 


--  /(m  +  1) 


(102) 


Thus  for  a  magnification  of  —  6  (inverted  image)  and  a  focal  length  of  0-5372  the  throw  is 
2.65jR.  In  the  reflective  optical  systems  contemplated  for  home  television  projection  re- 
ceivers R  =  13.7  in.,  and  so  for  a  magnification  of  —  6  the  throw  would  be  2.65  X  13.7  in. 
=  36.3  in. 

TUBE  FACE.  Before  striking  the  spherical  mirror,  the  light  emitted  by  the  lumi- 
nescent screen  of  the  cathode-ray  tube  first  passes  through  a  thickness  (generally  about 
1/8  in.)  of  glass  constituting  the  tube  face.  The  tube  face  should  preferably  consist  of  two 
concentric  surfaces,  and  thus  it  acts  as  a  weak  lens.  The  radius  of  curvature  of  the  outer 
surface  (for  a  thin  tube  face)  is  approximately  equal  to  the  focal  length  of  the  system. 
The  lens  action  of  the  tube  face  changes  the  magnification  of  the  system  to 


M 


-  i 


(103) 


N 


where  N,  t,  and  RI  are  the  index  of  refraction,  thickness,  and  radius  of  curvature  of  the 
outer  surface  of  the  tube  face.    However,  the  largest  effect  of  the  tube  face  is  caused  by 


14-26 


OPTICS 


images  cannot  be  seen.  The  blind  spot  is  roughly  elliptical,  the  vertical  dimension  being 
longer.  Its  size  varies  greatly  among  individuals,  with  limits  for  horizontal  dimensions 
of  3  to  8  degrees,  averaging  about  6  degrees.  The  nearest  edge  is  located  about  12  degrees 
on  the  nasal  side  of  each  retina. 

Contrary  to  other  sense  organs,  the  entire  nerve  system  leading  to  the  cortical  regions 
of  the  brain  is  found  essentially  in  the  retina  itself.  Here  some  of  the  functions  may  occur 
which  usually  take  place  in  the  cortex,  for  example  the  color-analyzing  processes  (Polyak). 

THE  REFRACTIVE  MEDIA.  The  transparent  intraocular  fluids  in  all  parts  of  the 
eye  are  derived  from  the  blood  and  physiologically  are  essentially  the  same.  The  fluid  in 


64 


72 


40 
Age,  Years 

FIG.  2.    The  Loss  of  Accommodation,  with  Age.    Data  from  over  4200  Eyes.    (Duane,  Am*  J.  Ophth.) 

the  anterior  chamber  (Fig.  1),  called  the  aqueous  humor,  is  only  slightly  more  viscous 
than  water,  is  subject  to  thermal  currents,  and  is  quickly  replenished  if  lost.  The  larger 
chamber  behind  the  crystalline  lens  is  filled  with  a  jellylike  substance  called  the  vitreous 
body.  This  body  is  a  combination  of  a  protein  colloid  and  the  intraocular  fluid,  also  per- 
meated by  a  fine  meshwork  of  fibrils  that  gives  the  mass  a  stable  anatomical  structure. 

The  crystalline  lens  is  a  transparent  body  having  the  shape  of  a  biconvex  lens,  and  it 
serves  the  same  function  as  a  lens.  It  consists  of  a  non-homogeneous  elastic  substance 
made  up  of  a  number  of  layers  or  laminae,  each  with  increasing  density  and  increasing 
index  of  refraction  toward  the  core  at  the  center.  By  the  action  of  the  ciliary  body,  through 
a  complex  process,  the  lens  may  become  more  convex  and  in  so  doing  serves  the  function 
of  changing  the  position  of  the  focal  point  of  the  light  entering  the  eye.  In  this  way,  the 
eye  can  accommodate  itself  from  distinct  vision  of  distant  objects  to  that  of  nearer  ob- 
jects, and  vice  versa.  In  this  act  of  accommodation  primarily  the  radius  of  the  front 
surface  decreases  and  the  thickness  of  the  lens  increases. 


THE  STRTJCTUBE  OF  THE  EYE 


14-27 


-7     -6    —5 


-3      -2      -1        O 
Log  Held  Luminosity,  ml 


With  increasing  age  the  average  density  of  the  lens  increases;  the  lens  becomes  harder 
and  less  elastic  and  hence  its  accommodative  function  is  less  and  less  effective.  The  loss 
of  accommodation  with  age  is  illustrated  in  Fig.  2  (Duane),  where  the  nearest  distance 
from  the  eye  at  which  a  test  object  remains  clear  is  plotted  against  age.  It  is  clear  thatf 
at  about  the  age  of  60  years,  little  or  no  accommodation  remains,  and  the  eye  is  then  said 
to  be  completely  presbyopic. 

THE  PUPIL.  The  pupil  acts  as  the  stop  or  aperture  of  the  eye.  It  is  in  a  constant 
state  of  activity  and  is  subject  to  a  number  of  reflexes.  Although  the  pupil  will  contract 
when  the  iris  is  stimulated  directly  by  light,  normally  it  contracts  through  a  light  stimula- 
tion of  the  retina,  the  extent  de- 
pending on  the  adaptation  of  the  8  f 
eye.  The  pupillary  response  to 
light  falling  on  the  central  part  of 
the  retina  is  much  greater  than 
for  that  falling  on  the  peripheral 
parts  of  the  retina.  The  thresh- 
old for  pupillary  response  of  the 
central  region  of  the  retina,  when 
dark  adapted,  corresponds  quite 
well  to  the  threshold  of  cone 
vision.  Data  showing  the  change 
in  the  diameter  of  the  pupil  in 
response  to  the  luminosity  of  an 
extended  surface  are  illustrated 
in  Fig.  3  (Reeves).  Theoretically 
the  amount  of  light  entering  the 
eye  would  be  approximately  pro- 
portional to  the  area  of  the  pupil 
(diameter  squared),  but  it  is  evi- 
dent from  the  figure  that,  whereas 
the  illumination  change  is  more  than  several  million  times,  the  amount  of  light  entering 
the  eye  changes  only  about  15  fold.  The  contraction  of  the  pupil  cannot,  therefore, 
even  approximately  [compensate  for  increasing  illumination.  The  rate  of  contraction  of 
the  pupil  from  darkness  to  light  is  much  greater  than  the  rate  of  dilation  resulting  from 
light  to  darkness.  The  response  of  the  pupil  to  colored  light  is  greater  for  yellow  than  for 
red  or  blue. 

In  this  connection,  it  has  been  shown  (Stiles  and  Crawford)  that  the  intensity  of  the 
light  falling  upon  the  retina  is  not  directly  proportional  to  the  pupil  area.  Rays  entering 

the  pupil  near  the  edge 

1-20, — , — i — | — , — | — i — , — | — , — i — j — i — , — , — i — , — j — , — , — ,     are  much   less  effective 

in  producing  a  sensation 
than  those  entering  the 
center  of  the  pupil.  The 
relative  effectiveness  of 
the  different  pupillary 
zones  is  shown  in  Fig.  4. 
The  pupil  of  the  eye 
contracts  when  the  eyes 
are  converged  and  ac- 
commodated for  near  ob- 
jects. This  assoeisiiQfi 
is  more  closely  related  to 
convergence  of  thte  eyes 
than  to  aca»mm<>datiQa. 
The  refer  takes  place  in. 
addition  to  efctaages  in 
pupilary  reaction  doe  to 
changes  in  illumination. 


FIG.  3.    The  Relation  between  Papillary  Si*e  and  the  Lu- 
minosity of  the  Visual  Field.    (Data  of  Ree-res,  J.  Optical  Sac. 
Am.) 


1.00 


0.80 


fcO.60 


0.20 


_S^ 


o  Horizontal  traverse' 
•  Verifca!  traverse     ; 


543210  12345 

Distance  from  Center  of  Pupil,  mm 

FIG.  4.    Data  Showing  the  Stiles-Crawford  Effect.    Light  through  periph-    „„„_„  ^ 

eral  zones  of  the  pupil  is  less  effective  than  that  through  the  center.    T~T^     +-„ 

(From  Moon,  Scientific  Bases  of  IUuminatinS  Engineering.)  Automatic 

of  the  pupil  for  near  ob- 

jects  increases  the  depth  of  focus  of  the  eye,  where  an  increased  depth  is  useful.  Under 
normal  conditions  pupillary  action  takes  place  nearly  equally  in  the  two  eyes,  even  if  only 
one  eye  is  subjected  to  iSumination  changes.  The  pupil  is  also  influenced  by  psychic 
factors;  a  dilation  is  usually  the  rule,  except  when  the  individual  is  in  a  comatose  state. 
There  is  some  evidence  that  pupillary  contraction  can  be  conditioned. 


14-28  OPTICS 

SCHEMATIC  EYE.  Because  of  the  non-homogeneity  of  the  crystalline  lens,  and  the 
general  asymmetry  of  the  optical  elements,  a  schematic  eye  can  be  only  an  approximation 
to  the  living  eye.  Frequently,  however,  such  a  schematic  eye  is  useful  for  reference  and 
for  optical  problems.  The  data  utilized  by  Gullstrand  for  the  relaxed  eye  are  given  in 
Table  1.  The  greater  part  of  the  refractive  power  of  the  eye  is  due  to  the  cornea  (43 
diopters),  the  lens  contributing  only  1/3  of  the  total  power.  Since  the  distance  of  the 
second  focal  point  from  the  second  nodal  point  of  the  schematized  eye  is  about  17  mm, 
an  object  subtending  an  angle  of  1  minute  of  arc  will  subtend  a  linear  distance  on  the  retina 
of  5  AC  (0.005  mm) .  In  problems  where  the  image  on  the  retina  must  be  considered  blurred, 
it  is  convenient  to  know  the  approximate  positions  of  the  entrance  and  exit  pupils.  All 
optical  imagery  calculations  can  be  referred  to  those  positions  if  the  magnification  of  the 
two  stops  are  known.  The  center  of  the  blurred  image  on  the  retina  may,  for  practical 
purposes,  be  taken  as  the  point  of  maximum  light  intensity. 

Table  1.    The  Gullstrand  Schematic  Eye  Relaxed  for  Distant  Vision 

Position  of  surfaces 

Cornea,  anterior  surface 0.0      mm 

Cornea,  posterior  surface 0.5 

Lens,  anterior  surface 3.6 

Core,  anterior  surface 4 . 15 

Core,  posterior  surface 6.56 

Lens,  posterior  surface 7.2 

Radii  of  curvature 

Cornea,  anterior  surface 7.7      mm 

Cornea,  posterior  surface 6.8 

Lens,  anterior  surface 10 . 0 

Core,  anterior  surface 7.91 

Core,  posterior  surface —  5 . 76 

Lens,  posterior  surface — 6.0 

Refractive  indices 

Cornea 1 .376  mm 

Aqueous  and  vitreous  humors 1 . 336 

Outer  portion  of  lens 1 . 386 

Core 1 .406 

Complete  Optical  System  of  the  Eye 

Position  of  first  principal  point 1 . 35    mm 

Second  principal  point 1 . 60 

First  nodal  point 7 . 08 

Second  nodal  point 7.33 

Anterior  focal  length — 17 . 05 

Posterior  focal  length 22 . 78 

Refractive  power  of  eye 58 . 64    diopters 

Position  of  entrance  pupil  from  cornea 2.0      mm 

Position  of  exit  pupil  from  posterior  surface  of  crystalline  lens 3.7 

Magnification  of  exit  to  entrance  pupils 0 . 923 

EYE  MOVEMENTS.  The  rotary  movements  of  each  eye  are  controlled  by  six  (ex- 
trinsic) muscles;  the  opposing  external  and  internal  recti  muscles  which  provide  move- 
ments for  looking  to  the  right  and  left;  the  superior  and  inferior  opposing  re cti-muscles 
which  provide  movements  for  looking  upwards  and  downwards;  and  the  two  oblique  mus- 
cles which  provide  torsional  movements  about  the  axes  of  fixation,  as  well  as  movements 
opposing  the  recti  in  certain  eye  positions.  The  innervations  to  the  muscles  for  movements 
of  the  two  eyes  are  said  to  be  reciprocal  in  that  a  given  movement  will  result  from  the  con- 
traction of  one  muscle  and  the  relaxation  of  its  antagonist.  There  is,  therefore,  a  precise 
coordination  of  the  muscles  that  leads  to  very  delicate  and  accurate  movements  of  the 
eyes.  In  general  it  can  be  said  that  the  ocular  movements  are  for  the  purpose  of  directing 
the  eyes  to  the  object  of  attention  and  preventing  diplopia  (double  vision).  The  move- 
ments seem,  essentially,  to  be  reflex  movements  following  the  direction  of  attention  and 
accordingly  have  been  called  psycho-optical  reflexes.  Because  the  actual  movements  of 
the  eyes  from  one  point  in  the  visual  field  to  another  appear  to  be  approximately  correct, 
it  has  been  assumed  that  the  retinal  elements  have  "motor  values,"  differences  in  which 
lead  to  correct  innervations  for  eye  movements.  The  reaction  time  between  the  atten- 
tion and  the  beginning  of  the  eye  movement,  though  varying  with  circumstances,  aver- 
ages between  0.17  and  0.20  sec. 

Owing  to  the  constant  tonus  of  the  muscles,  the  eyes  are  in  a  continuous  state  of  activity. 
Thus,  even  with  constant  fixation,  there  are  occasional  large,  jerky  movements  that  aver- 
age 4  minutes  of  arc  and  occur  at  intervals  of  1  to  2  sec.  In  between  these  are  smaller 
swinging  movements,  and  superimposed  on  both  are  very  small  vibratory  movements. 


THE   OPTICAL  CHARACTERISTICS  OF  THE   EYE       14-29 


^  The  voluntary  movements  are  said  to  be  conjugate  if  in  the  same  direction,  and  disjunc- 
tive if  in  the  opposite  direction,  as  when  the  eyes  converge  for  a  near  object.  The  limits 
within  which  the  eyes  can  move  without  head  movement  determine  the  field  of  fixation. 
Obviously,  this  will  vary  with  anatomical  features  of  the  head,  but  it  is  illustrated  typi- 
cally in  Fig.  5.  The  disjunctive  movements  are  primarily  concerned  with  convergence 
movements  within  which  binocular  fusion  can  be  maintained,  and  this  is  of  importance  in 
ophthalmology  since  abnormalties  in  this  function  often  lead  to  ocular  discomfort.  The 
"far  point"  of  convergence,  which  may  be  behind  the  head  (a  divergence  of  the  eyes),  can 
best  be  measured  by  ophthalmic  prisms,  while  the  "near  point"  can  be  measured  by 
bringing  a  small  object  nearer  and 
nearer  to  the  nose  until  doubling 
occurs.  Abnormalities  in  conver- 
gence are  frequently  associated 
with  refractive  errors  but  may  have 
more  deep-seated  innervational 
origins.  If  one  eye  is  covered 
while  the  other  fixates  a  given 
point,  the  covered  eye  may  deviate 
from  the  direction  of  the  point. 
This  deviation,  when  binocular 
vision  is  prevented,  is  called  hetero- 
phoria.  If  the  eye  deviates  out- 
ward, inward,  upward,  or  twists 
about  the  fixation  axis,  the  phorias 
are  said  to  be  exophoria,  esophoria, 
hyperphoria,  and  cyclophoria  re- 
spectively. Almost  everyone  ex- 
hibits at  least  a  small  phoria  for 
certain  visual  distances,  and  in 
fact  an  exophoria  of  about  2  to  3 
degrees  for  near  objects  would  be 
considered  normal.  Phorias  fre- 
quently are  indications  that  sus- 
tained efforts  are  being  made  to 
maintain  proper  convergence  of 
the  eyes  in  binocular  vision. 


Left  Eye 


SigHtEye 


FIG.  5.    An  Illustration  of  Typical  Monocular  and  Binocular 
Fields  of  Fixation  (from  Asher,  Arch.  f.  Opkthal.) 


In  reading  or  similar  visual  tasks,  the  eyes  make  a  series  of  short  interrupted  movements 
called  saccadic  movements.  Even  between  two  points,  the  movements  do  not  occur 
exactly  but  may  arrive  by  a  series  of  small  successive  approximations,  one  eye  usually 
leading.  Eye  movements  are  generally  quicker  in  the  horizontal  than  in  the  vertical 
directions,  and  lateral  movements  are  faster  than  convergence  movements.  Convergence 
movements  are  more  rapid  than  divergence  movements.  The  speed  of  movement  varies 
with  the  excursion,  attention,  and  other  conditions,  but  on  the  average  is  between  100 
and  200  degrees  per  second. 


8.  THE  OPTICAL  CHARACTERISTICS  OF  THE  EYE 

ABERRATIONS.  In  addition  to  the  usual  aberrations,  the  image  on  the  retina  of 
the  eye  suffers  also  from  irregular  defects  due  to  non-homogeneities  in  structure  and  the 
lack  of  symmetry  in  the  optical  media.  The  axis  of  the  crystalline  lens  is  tipped  and  de- 
centered  with  respect  to  that  of  the  cornea.  The  visual  axis,  which  is  determined  by 
the  fovea  as  well  as  the  pupil,  is  usually  decentered  with  respect  to  both  the  axes  of  the 
cornea  and  the  lens.  The  spherical  aberration  is  asymmetrical  From  tbe  central  zone 
of  the  pupil  to  the  outside  zones  the  eye  tends  to  be  myopic,  increasing  at  first  rapidly, 
then  reaching  a  maximum  at  1  mm,  beyond  which  the  myopia  decreases  slowly,  Tbe 
problem  of  the  spherical  aberration  of  the  eye  is  a  complex  one,  and  there  are  not  yet 
enough  data  to  understand  it  satisfactorily.  The  chromatic  aberration  of  the  eye  from 
the  C  (656  mji)  to  the  F  (487  m/*)  lines  of  the  hydrogen  spectrum  amounts  to  about  0.7 
diopter.  The  red  rays  are  focused  to  a  point  beyond  the  retina;  blue  rays,  to  a  point  ante- 
rior to  the  retina.  The  eye  is  said  to  be  hyperopic  to  red,  myopic  to  blue  light.  The 
change  in  chromatic  foci  for  equal  changes  in  wavelength  is  much  less  toward  the  red  but 
increases  markedly  toward  the  blue.  No  entirely  satisfactory  explanation  has  been  given 
as  to  why  chromatic  halos  about  images  of  point  light  sources  are  not  seen  by  tbe  eye.  For 
emmetropes  (individuals  with  no  refractive  error)  the  eye  focuses  approximately  for  the 
D  (580  mp)  or  yellow  line  of  hydrogen. 


14-30  OPTICS 

Astigmatism  at  oblique  incidence  is  a  larger  aberration.  The  primary  (tangential) 
astigmatic  field  differs  only  slightly  from  the  surface  of  the  retina  out  to  20  degrees,  while 
the  secondary  field  rapidly  increases  in  the  myopic  sense  to  nearly  11/4  diopters  at  a 
peripheral  angle  of  20  degrees.  There  is  very  little  coma  in  the  eye,  and  the  curvature  of 
the  field  approximates  the  curvature  of  the  retina  itself.  The  retinal  image  is  believed  to 
have  a  slight  barrel  distortion,  though  this  is  difficult  to  determine  because  of  the  subjec- 
tive asymmetries  of  the  retinas. 

These  aberrations  in  themselves,  together  with  diffraction  phenomena,  would  tend  to 
reduce  greatly  the  resolving  power  of  the  eye.  However,  as  Gullstrand  has  pointed  out, 
these  aberrations  together  produce  a  complex  caustic  surface  in  the  cone  of  light  converg- 
ing toward  the  retina,  so  that  actually  the  area  of  maximum  light  concentration  in  the 
image  may  be  small.  This  makes  possible  a  resolving  power  whose  limit  is  determined 
only  by  the  diameter  of  the  receptor  nerve  endings. 

REFRACTIVE  ERRORS.  The  eye  can  be  defective  as  the  result  of  anomalies  in  the 
refractive  media  or  in  the  length  of  the  eyeball,  when  it  is  said  to  be  ametropic.  The  result 
is  blurred  retinal  imagery,  with  the  attendant  loss  in  sharpness  of  vision.  Regular  ame- 
tropia  is  of  two  types — spherical  and  astigmatic.  The  first  results  from  symmetrical 
abnormalities  of  the  refractive  media,  or  from  an  increased  or  decreased  length  of  the  eye- 
ball. Astigmatic  errors  result  only  from  non-sphericity  of  the  surfaces  of  the  cornea 
and/or  of  the  crystalline  lens.  When  the  image  focuses  inside  the  position  of  the  retina, 
the  eye  will  be  nearsighted  (myopic),  because  only  when  the  objects  are  brought  to  the 
eye  will  the  images  move  back  to  the  retina  and  appear  sharply  defined.  In  the  reverse 
case,  when  the  point  of  focus  is  behind  the  retina,  the  eye  is  said  to  be  farsighted  (hyper- 
opic),  because  nearer  objects  are  the  more  blurred.  Nearsightedness  can  be  corrected 
with  a  minus  (diverging)  ophthalmic  lens  of  proper  power  before  the  eye;  farsightedness 
with  a  positive  (converging)  lens.  When  astigmatism  is  present,  the  light,  after  entering 
the  eye,  focuses  not  as  a  point  but  as  two  lines  separated  in  space  and  at  right  angles  to 
each  other.  Images  of  lines  in  space  which  are  parallel  to  that  focal  line  nearest  the  retina 
will  be  seen  more  distinctly  than  those  lines  at  right  angles.  For  astigmatism,  the  corrective 
ophthalmic  lens  before  the  eye  must  have  one  toric  surface.  Astigmatism  may  (and  usually 
does)  occur  in  combination  with  spherical  refractive  errors. 

Usually,  when  a  refractive  error  is  corrected  by  an  ophthalmic  lens,  the  magnification 
of  the  retinal  image  cannot  be  predicted.  A  spherical  refractive  error  due  to  an  elongation 
(or  shortening)  of  the  eyeball,  when  corrected  by  a  lens,  results  in  practically  no  magnifica- 
tion (or  diminution)  of  tbe  retinal  images.  If  the  refractive  error  is  due  to  abnormalities 
in  the  refractive  media,  then  a  magnification  (if  the  correcting  lens  is  for  farsightedness)  or 
a  diminution  (if  the  correcting  lens  is  for  nearsightedness)  will  occur.  The  degree  of 
magnification  (or  diminution)  will  be  roughly  1  1/2  to  2  per  cent  per  diopter  power  of  the 
correcting  lens.  Differences  in  the  magnification  of  the  images  of  the  two  eyes  may  result 
in  discomfort. 

DEPTH  OF  FOCUS.  As  in  any  optical  instrument,  the  image  falling  upon  the  retina 
of  the  eye  can  be  slightly  blurred,  because  of  the  eye's  being  out  of  focus,  without  an  appre- 
ciable loss  of  perceived  definition  of  the  image.  Thus,  with  the  eye  accommodated  (fo- 
cused) for  a  given  distance,  objects  somewhat  nearer  and  farther  than  this  distance  will 
appear  clear.  The  distance  between  the  two  limits  of  visual  distance  within  which  the 
images  on  the  retina  appear  not  to  suffer  loss  of  definition  is  called  the  depth  of  focus. 
This  fact  arises,  in  part,  because  the  maximum  light  concentration  is  in  the  center  of  the 
blur  circle.  Only  when  the  intensity  of  this  central  portion  is  reduced  as  the  blur  circle 
is  broadened  will  there  be  a  loss  in  definition.  The  spherical  and  other  aberrations  of  the 
eye  are  such  that  the  depth  of  focus  is  greater  than  might  be  anticipated  from  ordinary 
theoretical  expectations.  Owing  to  the  constriction  of  the  pupil  with  accommodation  and 
convergence,  the  depth  of  focus  increases  with  the  nearness  of  the  fixation  object.  For 
distance  vision  the  depth  of  focus,  measured  as  the  difference  of  the  reciprocals  of  the 
nearer  and  farther  limiting  distances  (in  meters)  for  clear  vision,  is  about  0.3  diopter, 
which  increases  to  about  0.7  diopter  for  a  visual  distance  of  20  cm.  With  the  eyes  relaxed 
for  distant  vision,  and  taking  into  account  that  the  conjugate  point  to  the  retina  is  usually 
not  at  infinity,  one  could  expect  all  objects  beyond  12  ft  to  be  clearly  defined.  With 
the  eyes  accommodated  for  the  reading  distance  of  16  in.  from  the  eyes,  all  objects  from 
about  14  1/2  to  17  !/2  in.  will  appear  clearly  defined. 

9.  THE  LIGHT  SENSE 

THRESHOLD  OF  LIGHT  VISIBILITY.  It  must  be  clear  that  all  determinations  of 
visual  thresholds,  involve  problems  in  psy chometrics  (Guilf  ord) .  A  given  weak  stimulus 


THE  LIGHT  SENSE 


14-31 


may  not  always  result  in  a  definite  response,  and  a  still  weaker  stimulus  will  be  responded 
to  even  less  often.  One  determines,  then,  the  percentage  of  responses  for  a  number  of 
exposures  to  the  same  stimulus  strength.  The  stimulus  strength  is  then  varied  in  fixed 
steps.  What  percentage  of  responses  corresponds  to  the  threshold  is  a  matter  of  defini- 
tion and  varies  with  the  experimenter.  It  is  well  to  know  this  in  trying  to  understand 
and  compare  threshold  data. 

As  a  device  for  detecting  radiant  energy,  the  eye  is  exceedingly  sensitive,  though  its 
response  is  confined  to  a  narrow  region  of  the  spectrum,  from  about  360  mp  to  75O  m/i. 
The  threshold  of  perception  of  light  varies,  of  course,  with  the  wavelength  of  the  light,  the 
size  of  the  stimulus,  and  the  length  of  exposure  to  the  stimulus.  It  is  also  different  for 
the  cone  and  rod  systems  of  the  retina.  Under  optimum  conditions,  and  taking  the 
threshold  as  the  60  per  cent  response  to  exposures,  Hecht  found  that  with  different  ob- 
servers the  threshold  varied  between  2.2  and  5.7  X  1CT18  erg  measured  at  the  cornea, 
which  amounts  to  about  58  to  148  quanta  of  light  energy-.  In  this  experiment  a  test  light 


was  used,  which  subtended  a  visual  angle  of  10  minutes 


stimulus  of  wavelength  510 
of  arc,  and  which  was  exposed 
as  a  flash  of  0.001-sec  duration. 
The  test  light  was  arranged  to 
stimulate  rod  vision  20  degrees 
temporally  from  the  fovea,  and 
the  subject  was,  of  course, 
completely  dark  adapted. 

The  relative  thresholds  of 
visibility  for  complete  dark 
adaptation  of  the  rods  and 
cones  of  the  fovea  for  different 
wavelengths  are  illustrated  in 
Fig.  6  (Wald).  Here  the  spec- 
tral sensitivities  (reciprocal  of 
the  'threshold  stimulus 
strength)  are  plotted.  The 
stimulus  was  a  circular  field 
subtending  a  visual  angle  of  1 
degree,  and  this  was  exposed 
for  1/25  sec.  For  the  r»d  vision 
a  point  8  degrees  above  the 
fovea  was  used.  It  is  clear 
from  the  figure  that  the  rods 
at  an  angle  of  8  degrees  into 
the  periphery  of  the  retina  are 
2.5  times  more  sensitive  than 

the  cones  at  the  fovea.  The  maximum  sensitivity  (lowest  threshold)  for  the  cones  occurs 
at  562  mju  whereas  that  for  the  rods  occurs  at  about  505  m./*.  The  displacement  of  the 
maxima  toward  the  blue  end  of  the  spectrum  from  cone  to  rod  vision  is  known  as  tiie 
Purkinje  phenomenon.  The  fact  that  at  the  red  end  of  the  spectrum  the  rods  and  coaes 
have  essentially  the  same  threshold  sensitivities  is  also  of  special  importance. 

The  exact  relationship  between  the  area  of  the  light  stimulus  and  its  intensity  just  at 
the  threshold  of  visibility  is  not  entirely  known.  For  the  fovea,  with  areas  less  tbaa  10 
minutes  of  arc,  and  for  the  periphery,  for  areas  between  2  and  7  degrees,  the  product  ol 
the  area  and  intensity  is  approximately  a  constant.  This  fact  is  known  as  Rieco's  law  for 
the  fovea  and  Piper's  law  for  the  periphery.  These  laws  are  wholly  inadequate  for  wider 
ranges  of  area,  and,  in  general,  for  larger  areas  the  product  of  area  and  intensity  appears 
to  be  a  decreasing  function  of  area. 

In  a  somewhat  similar  manner,  with  a  constant  area,  the  relationship  between  intensity 
and  the  duration  of  the  stimulus  is  not  fully  understood.  For  brief  flashes  of  duration  ol 
less  than  0.2  sec,  and  with  grnnTO  areas,  the  product  of  duration  of  the  flash  and  the  intensity 
is  approximately  constant  (Bloch's  law).  However,  for  longer  durations  this  constancy 
no  longer  holds,  and  finally  the  intensity  alone  becomes  the  determining  criterion  for 
threshold  perception.  It  has  also  been  found  that  the  stimulation  of  one  part  of  toe  retina 
by  a  small  light  source  depresses  the  sensitivity  of  other  parts  of  t3be  retina  for  simultar- 
neous  but  not  for  succeeding  stimuli. 

SPECTRAL  LTTMIlf  OSITTES,  The  so-called  visibility  curves  for  day  light  (yhoto&c) 
vision  and  twilight  (scofopic)  vision  are  obtained  in  brightness-matcMng  experiments-  A 
small  area  illuminated  by  a  narrow  region  of  the  daylight  spectrum  is  presented  adjacent 
to  a  standard  field  of  constant  brightness  and  one  to  which  the  eye  is  adapted.  The  sub- 


FIG.  6.    Spectral  Senativities  of  the  Rods  and  Cones  of  the  Eye, 

Expressed  Relative  to  the  Maximum  Sensitivity  af  the  Fovea 

(Wald,  Science) 


14-32 


OPTICS 


0.8 


0.7 


0.6 


0.5 


0.3 


0.2 


0,1 


A  Aubert 
o  Koenlg 
•  Brodhun 
ABIancha.rd 


-5      -4 


—3       -2-1          0          1 
Log  LuminosIty-MIIIIIamberts 


PIG.  7.    Typical  Data  Showing  the  Differential  Threshold  of 
Brightness  Discrimination  (from  Hecht,  J.  Gen.  PhysioL) 


ject  adjusts  the  intensity  of  the  colored  light  in  the  test  area  so  that  it  is  apparently  equal 

to  the  brilliance  of  the  standard 
field.  The  reciprocal  of  the  inten- 
sity of  the  transmitted  spectral 
colors  for  the  match  measures  the 
relative  brightness  values  of  the 
visible  spectrum.  The  data  are  ob- 
tained when  the  brightness  of  the 
standard  field  and  the  surround- 
ings correspond  to  average  day- 
light intensities  and  again  to  twi- 
light visual  conditions.  Figure  15, 
p.  14-14,  shows  the  curves  which 
are  usually  found  but  in  which  the 
maxima  are  adjusted  for  the  same 
height.  With  luminosities  above 
about  0.01  lumen  per  sq  ft,  the 
photopic  or  cone  visibility  curve, 
which  has  a  maximum  at  about 
555  m/u,  is  found,  while  with  lumi- 
nosities below  about  0.001  lumen 
per  sq  ft,  the  scotopic  or  rod 
visibility  curve  is  found,  with  a 
maximum  at  about  507  mju.  For 
brightness  levels  between  0.01  and 
0.001  lumen  per  sq  ft  there  is  a 
gradual  shift  of  the  position  of  the 
curve,  this  being  a  transition  from 
cone  to  rod  vision.  The  displace- 
ment of  the  maximum  is  again  the 
Purkinje  phenomenon.  In  very 
low  illuminations  the  eye  responds  less  and  less  to  the  red  while  still  responding  to  the 
blue.  No  PurMnje  displace- 
ment is  found  for  foveal  vision 
alone. 

DIFFERENTIAL  SENSI- 
TIVITY. The  smallest  differ- 
ence in  brightness  that  can 
just  be  perceived  (just-notice- 
able difference)  between  two 
areas  is  a  measure  of  the 
differential  threshold  of  visi- 
bility from  which  one  arrives 
at  a  value  of  the  differential 
sensitivity,  or  contrast  sensi- 
tivity. The  differential  thresh- 
old varies  greatly  with  the 
illumination,  the  wavelength, 
the  size  and  separation  of  the 
test  areas,  and  the  luminosity 
of  the  surroundings.  The 
differential  threshold  is  ex- 
pressed as  the  ratio  AJ/I, 
where  1-  is  the  luminosity  of 
the  standard  area  and  the 
^adapting  field,  and  AJ  is  the 
difference  in  luminosity  of  the 
test  area  and  the  standard 
area  for  the  just-noticeable 
difference  in  brightness.  The 
contrast  sensitivity  is  the 
reciprocal  of  this  quantity. 
Typical  differential  threshold 


Contcast  Sensitivity  •  I/Al 

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FIG.  8.    The  Contrast  Sensitivity  of  the  Eye,  Showing  Data  of 

Koenig,  Blanchard,  Holliday,   and   Stiles  and  Crawford  (after 

Moon,  Scientific  Bases  of  Illuminating  Engineering) 


results  are  shown  in  Fig.  7,  showing  data  recomputed  by  Hecht.    Because  of  the  wide  range 
in  luminosities  necessary,  the  abscissas  are  plotted  on  a  logarithmic  scale.    In  the  photopic 


TEMPORAL  ASPECTS  OF  PERCEPTION 


14-33 


part  of  the  curve  the  fraction  AJ/7  is  nearly  constant,  and  only  in  this  region  can  the 
differential  threshold  of  the  eye  be  said  to  illustrate  Weber's  law.  For  very  high  luminosi- 
ties a  slightly  decreased  sensitivity  may  be  found,  but  this  can  be  attributed  to  uncon- 
trolled illumination  of  the  more  peripheral  surroundings  of  the  test  areas.  That  part  of 
the  curve  below  0.013  millilambert  results  from  rod  vision  only,  and  a  somewhat  abrupt 
change  near  that  point  is  suggested  by  the  data.  It  is  sometimes  more  instructive  to  plot 
the  differential  threshold  data  as  the  contrast  sensitivity  <//AZ),  as  shown  in  Fig.  8. 

If  smaller  test  and  standard  areas  are  used,  the  sensitivity  is  less,  that  is,  the  curve  lies 
higher  on  the  AJ/7  axis  of  the  differential  threshold  graph.  Also  it  is  found  that,  if  the 
luminosity  of  the  surroundings  outside  of  the  test  and  standard  areas  is  kept  constant, 
and  then  that  of  the  standard  area  is  varied,  the  differential  threshold  is  increased  (sensi- 
tivity decreased),  and  the  entire  curve  lies  above  the  curve  shown  in  Fig.  7.  The  luminos- 
ity of  the  surroundings  of  the  test  field  is  an  important  factor  in  the  contrast  sensitivity 
of  the  eye. 

ADAPTATION.  When  one  goes  from  a  brightly  illuminated  room  to  one  that  is  only 
dimly  illuminated,  several  minutes  must  elapse  before  details  in  the  room  can  be  discerned. 
Likewise  in  going  from  the  dark  room  into  sunlight  there  are  a  few  moments  of  blinding 
glare.  In  either  case,  the  eyes  become  adjusted  to  the  prevailing  brightness  of  the  vitual 
field  in  a  few  minutes.  The  retinal  process  by  which  this  occurs,  as  well  as  the  firm!  sta- 
tionary state,  is  called  adaptation. 
By  the  process  of  adaptation  the 
sensitivity  of  the  eye  to  contrast 
differences  is  automatically 
changed  to  meet  changing  lumi- 
nosities of  the  surroundings.  In 
light  adaptation  (photopic  vision) 
the  visual  sensitivity  of  the  eye  de- 
creases;  in  dark  adaptation  (sco-  J; 
topic  vision)  it  increases.  In  this 
adaptation  the  eye  adapts  itself  to 
changes  of  illumination  of  several 
thousand  times. 

Light  and  dark  adaptations  are 
opposing  processes  and  are  differ- 
ent for  the  cone  and  rod  systems 
of  the  retina.  To  a  small  extent 
the  pupil  aids  in  the  adaptation 
processes.  A  typical  curve  show- 
ing the  progress  of  dark  adaptation  of  the  eye  as  a  whole  is  shown  in  Fig.  9.  The  ordinates 
indicate  the  luminosity  of  the  test  object  at  the  threshold  of  visibility  at  any  gi\~en  mo- 
ment in  the  progress  of  the  dark  adaptation.  The  upper  curve  results  from  the  activity 
of  the  retinal  cones;  the  remaining  part  of  the  curve  results  from  the  rods.  The  data  show 
that  foveal  (cone)  dark  adaptation  is  nearly  complete  in  2  to  10  min.  For  the  rods  the 
eye  is  essentially  dark  adapted  in  40  min,  though  the  sensitivity  can  still  be  shown  to  in- 
crease slightly  for  longer  periods  of  time  in  total  darkness.  Anomalies  in  dark  adaptation 
may  be  due  to  vitamin  A  deficiency. 

The  progress  of  dark  adaptation  varies  somewhat  with  the  siae  of  the  test  object  and 
especially  with  the  intensity  of  the  illumination  in  the  previous  light  adaptation.  There 
is  evidence  also  that  the  rate  of  dark  adaptation  can  be  increased  and  the  dark  adaptation 
maintained  when  the  observer  wears  dark  red  glasses,  adequately  shielded,  in  ordinary 
illuminations  (Rowland  and  Sloan). 

The  rate  of  light  adaptation  is  very  much  more  rapid  than  dark  adaptation.  Though 
dependent  upon  the  intensity  of  the  adapting  light,  the  sensitivity  drops  to  a  fraction  of 
its  initial  value  within  the  first  few  seconds,  and  the  light  adaptation  is  nearly  complete 
in  1  min. 

10.  TEMPORAL  ASPECTS  OF  PERCEPTION 

PERSISTENCY  OF  VISION.  From  the  instant  the  retina  is  stimulated  to  the  moment 
the  first  sensation  occurs,  a  brief  interval  of  time  has  elapsed.  This  interval,  called  the 
latent  period,  which,  in  a  sense,  is  a  visual  reaction  time,  depends  upon  the  nature  and 
strength  of  the  light  stimulus  and  upon  the  adaptation  of  the  retina.  Under  ordinary 
light  conditions,  the  latent  period  is  roughly  between  0.06  and  0.2  sec.  As  the  intensity 
of  the  stimulus  is  increased,  however,  the  latent  period  decreases  to  a  minimum  (0.065  to 
0.130  sec),  beyond  which  further  increases  in  intensity  will  not  shorten  the  latent  period. 


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FIG.  9.    The  Dark-adaptation  Curve  of  the  Eye  as  a  Whole 

and  of  Lhe  Fovea  Showing  the  Behavior  of  the  Cone  and  Rod 

Svstems   (Hecht,  J.  Gen.   Pkysiol.) 


14-34 


OPTICS 


Above  the  minimum,  this  interval  varies  approximately  inversely  as  the  logarithm  of  the 
intensity.  In  dark  adaptation,  the  latent  period,  for  threshold  conditions,  may  be  0.5  to 
1  sec,  depending  upon  conditions.  The  latent  period  is  generally  shorter  in  the  peripheral 
parts  of  the  retina  than  at  the  fovea,  and  shorter  for  blue  light  than  for  red. 

With  a  flash  stimulus,  the  sensation  persists  for  some  time  after  the  stimulus  has  ended. 
The  magnitude  of  the  light  sensation  and  its  duration  vary  with  the  adaptation  level  of 

the  eye,  that  is,  the  luminosity  of  the 

8 1 1 1 j \ 1 1 1     background  upon  which  the  stimulus 

is  seen  and  the  strength  of  the  stimu- 
lus. Typical  curves  are  shown  in 
Fig.  10.  The  duration  of  the  im- 
pressions from  stimuli  of  equal 
brightness  is  longer  for  the  fovea 
than  for  the  peripheral  parts  of  the 
retina  and  is  generally  increased  as 
the  eye  is  more  dark  adapted.  An 
increased  intensity  of  the  stimulus 
results  in  a  decreased  duration  time. 
For  colored  light  the  persistence  of 
the  sensation  varies  inversely  to  the 
apparent  luminosities  of  the  visibil- 
ity curves  and  hence  is  shortest  for 
yellow,  longest  for  blue,  and  inter- 
mediate for  red. 

After  the  first  sensation  has  faded 
away,  several  after-images  may  again 
appear  at  intervals  depending  upon 
the  luminosity  of  the  surroundings 
and  the  intensity  of  the  stimulus. 
If  the  after-image  corresponds  to  the 
original  impression  as  regards  con- 
trast and  color  it  is  called  positive; 
S?G*  I?'  ffi^teatiOT  of  the  Apparent  Brightness  and  the  jf  the  Hght-to-dark  relationships  are 
Duration  of  the  Light  Sensation  Following  a  Flash  Stun-  f  ,  , ,  ,  i 

ulus  (Data  of  Broca  and  Sulzer,  from  Luckiesh  and  Moss,     reversed  and  the  colors  are  comple- 
Stience  of  Seeing)  mentary  to  the  original  impression 

it  is  called  negative.     Considerable 

practice  is  sometimes  necessary  to  see  these  images,  for  conditions  of  fatigue,  adaptation, 
and  inhibition  may  greatly  affect  their  appearance.  The  typical  sequence  of  after-images 
is  illustrated  in  Fig.  11,  where  the  ordinate  represents  the  relative  brightness  of  the  after- 
images, positive  above  and  negative  below  the  abscissa. 


0.2 

Time  In  Seconds 


VIII  (6) 


667 
Time  In  Seconds 


10 


11 


FIG.  11.    Diagram  of  the  Sequence  of  After-effects  Folio-wing  a  Flash  Stimulus  in  the  Light-adapted  Eye 
(after  Tschermak-Bethe,  Handb.  normal  u.  path.  Physiol.,  12/1,  II) 

PERIODIC  STIMTJLI  AND  FLICKER.  Periodic  light  stimuli  may  be  perceived  as 
discrete  flashes,  but  if  made  sufliciently  rapid,  so  that  the  persistent  sensation  from  one 
stimulation  curve  overlaps  the  rise  of  the  primary  sensation  from  the  succeeding  stimula- 
tion, the  sensation  will  be  the  same  as  that  for  a  continuous  illumination. 

The  brilliance  of  fused  periodic  stimuli  which  have  different  intensities  is  the  same  as 
the  average  intensity  (TaJbot's  law),  that  is,  the  integral  of  the  photometric  luminosity 
of  the  periodic  stimuli  divided  by  tune.  This  law  holds  accurately  except  for  extremes  of 
high  and  low  intensities.  That  frequency  at  which  periodic  stimuli  are  first  perceived  as 
a  steady  illumination  (fused)  is  known  as  the  critical  flicker  frequency,  which  may  be 
abbreviated  to  e.f .f.  The  critical  flicker  frequency  varies  with  the  illumination,  the  part 


COLOR 


14-35 


of  retina  being  stimulated,  the  area  of  the  flickering  field,  the  ratio  of  the  light-to-dark 
intervals  of  the  flashes,  the  retinal  adaptation,  the  wavelength  of  the  light,  and  the  pres- 
ence of  other  steady  light  stimuli  also  falling  upon  the  retina.  So  reliable  is  the  c.fX 
under  controlled  conditions  that  it  is  sometimes  used  to  measure  luminosities  and  adapta- 
tion levels. 

The  manner  in  which  the  c.f .f .  for  small  test  fields  illuminated  with  white  light  varies 
with  illumination  and  for  different  parts  of  the  retina  is  shown  in  Fig.  12.  In  the  rod-free 
area  at  the  fovea,  the  c.f.f.  varies  proportionately  with  the  logarithm  of  the  illumination 
(Ferry-Porter  law),  except  at  the  extremes  of  the  iUuminatioa.  The  maximum  critical 
frequency  is  about  53  cycles  per  second.  The  c.f.f.  decreases  for  flicker  stimuli  on  the 
peripheral  parts  of  the  retina.  If  the  area  of  the  flickering  field  is  large,  however,  the 


60 


W 

o 


20° 


-5 


—3 


-1 


FIG.  12.    The  Relationship  between  the  Critical  Flicker  Frequency  and  the  Illumination,  for  Different 
Parts  of  the  Retina  (Hecht,  J.  Gen.  P&yswjJ.) 

critical  frequency  may  be  found  higher  in  the  periphery,  which  indicates  a  spatial  summa- 
tion of  the  retina  in  the  periphery.  The  curves  generally  appear  to  be  in  two  parts,  a 
division  which  is  attributed  to  the  rod  and  cone  behavior.  The  c.f.f.  for  various  colored 
stimuli  is  substantially  the  same  as  for  white  light  if  their  apparent  luminosity  is  the  same 
as  that  of  the  white  light  and  the  intensity  is  above  the  cone  threshold. 

In  the  curves  shown  in  Fig.  12,  the  duration  of  the  light  phase  was  equal  to  that  of  the 
dark.  Where  the  dark  phase  is  longer,  generally  higher  critical  frequencies  are  found,  and 
when  shorter,  lower  critical  frequencies  are  necessary,  especially  in  the  ordinary  ranges,  of 
illuminations.  Only  at  high  intensities  of  the  light  stimuli  will  this  generalization  be  in- 
valid. Rapid  eye  movements  or  rapidly  moving  periodic  stimuli  enhance  the  appearance 
of  flicker,  and  under  these  conditions  higher  frequencies  are  necessary.  A  steady  illumina- 
tion of  one  part  of  the  retina  changes  the  c.f.f.  of  a  periodic  stimulus  in  another  part  if 
the  separation  of  the  two  stimuli  is  not  too  great.  In  general  the  c.f.f.  increases  as  the 
brightness  of  the  steady  stimulus  is  increased,  until  a  maximum  is  reached,  beyond  which 
it  then  decreases. 

Hartley  found  that  for  periodic  stimuli  below  the  c,f.f.,  and  at  about  8-10  flashes  per 
second,  there  is  a  marked  enhancement  of  the  apparent  brightness  of  the  stimuli. 


11.  COLOR 

A  clear  differentiation  must  be  made  between  the  color  s*w»«Ius  and  the  color  sensation. 
The  color  sensation  and  all  the  associated  phenomena  are  purely  psychological  (experien- 
tial). The  stimuli  measured  in  terms  of  energy  and  wavelength  are  physical.  The  color 
problem  is  fundamentally  the  finding  of  the  relationships  that  exist  between  the  color 


14-36 


OPTICS 


sensation  and  the  physical  stimuli.  Moreover,  relationships  found  for  color  mixtures 
from  spectral  light  will  not  necessarily  be  true  when  colored  pigments  are  mixed.  This 
fact  has  led  to  much  confusion  and  controversy.  The  spectral  luminosities  reflected  from 
pigments,  however,  are  subject  to  the  same  laws  as  those  from  spectral  light. 

Colors  are  ordinarily  seen  as  properties  of  objects  and  hence  are  usually  associated  with 
the  objects  themselves.  There  are,  therefore,  many  psychological  constancy  phenomena 
in  which  objects  tend  to  retain  their  color  and  brightness  when  seen  in  varying  conditions 
of  light  and  shade.  Photographic  scenes  in  color  projected  in  an  otherwise  darkened  room 
tend  to  retain  normal  color  relationships  in  spite  of  actual  wide  deviations.  Under  other 
conditions  where  these  reproductions  are  viewed  against  backgrounds  of  stable  color  and 
contrast  relationships,  these  deviations  are  more  readily  apparent.  To  study  correlations 
between  the  quality  of  color  experience  and  the  physical  composition  of  light,  the  test 
fields  must  be  dissociated  from  known  objects  having  spatial  values.  These  psychological 
facts  are  important  in  dealing  with  color  problems. 

A  given  color  sensation  is  said  to  have  three  dimensions:  hue,  saturation,  and  brilliance. 
Hue  is  associated  with  the  dominant  wavelength  of  the  light  stimulus.  Saturation  indi- 
cates the  amount  or  degree  of  the  hue  present  (deep-red  as  against  pale-red)  and  repre- 
sents the  ratio  of  the  luminosity  of  the  pure  spectral  light  to  that  of  the  white  light  present. 
Brilliance  (apparent  brightness,  apparent  luminosity)  indicates  the  total  intensity  (energy) 
of  the  colored  light  stimulus.  Frequently,  hue  and  saturation  are  considered  together 
under  the  term  chromaticity, 

SATURATION  OF  THE  SPECTRAL  COLORS.  All  spectral  colors  do  not  appear 
equally  saturated,  that  is,  some  have  a  greater  sensation  of  white  than  others.  The  ratio 
of  the  luminosity  of  the  least  perceptible  spectral  color  to  the  luminosity  of  a  background 
field  defines  the  least-perceptible  colorimetric  purity  for  that  color.  Two  halves  of  a  test 
field  are  equally  illuminated  with  white  light.  To  one  half  a  spectral  color  is  added,  and, 
in  order  that  the  brightness  of  the  two  fields  shall  be  the  same,  the  luminosity  of  the  white 
light  of  that  field  is  decreased  as  the  spectral  color  is  added.  The  intensity  of  this  spectral 
color  is  then  increased  until  the  two  halves  of  the  field  just  appear  different  in  color.  From 
the  luminosity  of  the  added  spectral  colors  the  colorimetric  purity  of  the  spectrum  can  be 
found.  Figure  13  illustrates  data  obtained  in  this  manner.  It  is  clear  that  a  great  deal 


O.ioo 


0.050 


0.001 


400 


500 


Wavelength-m^t 


600 


700 


FIG.  13.    The  Least-perceptible  Colorimetric  Purity  of  the  Visible  Spectrum  (Data  of  Priest  and  Brick- 
wedde,  after  Hecht,  J.  Gen.  Physiol.} 

more  of  yellow  than  red  or  blue  is  required  to  produce  a  least-perceptible  color,  and  yellow 
is  therefore  considered  a  less  saturated  color  than  red  or  blue. 

HTTE  DISCRIMINATION  THRESHOLD.     The  change  in  wavelength  corresponding 
to  a  just-noticeable  difference  in  color  varies  over  the  spectrum  and  also  somewhat  with 


COLOR 


14-37 


the  individual.    In  general,  however,  the  data  are  similar,  and  the  resultant  curves  show 
three  maxima  and  three  minima.    In  Fig.  14  are  illustrated  representative  data  for  an 
individual  with  normal  color  vision.   The  dis- 
crimination is  poorest  at  the  ends  of  the  spec- 
trum, especially  in  the  red. 

COLOR  SPECIFICATION.  The  color 
stimulus  (as  distinct  from  the  color  sensation) 
of  any  color  field  can  be  specified  quantita- 
tively by  the  luminosity  of  the  radiation  given 
off  for  each  part  of  the  spectrum.  This  can 
usually  be  measured  by  a  speetroradiometer 
or  a  spectrophotometer.  Under  any  condi- 
tion, the  objective  color  from  a  given  surface 
can  then  be  specified  by  a  spectrophotometric 
curve  obtained  from  those  measurements. 

The  eye  itself  cannot  analyze  the  radiation 
from  a  color  stimulus  as  can  the  spectro- 
photometer, for  it  responds  in  a  complex 
manner  dependent  not  only  upon  the  visual 
processes  and  their  reaction  to  light  stimuli 
but  also  upon  certain  psychological  factors. 
There  are,  therefore,  many  different  objective 
color  stimuli,  as  specified  by  spectrophoto- 
metric curves,  which  will  result  in  the  same 
color  sensation.  It  is  possible,  however,  to 
transform  the  data  from  the  spectrophoto- 


4OO  440   4SO   52O  56O  600   640  680 


FIG.    14.     The    Least-pereeptibk   IKfferasee  in 

Color  for  the  Viable  Spectrum  (Data  of  Jones, 

J.  Optical  See.  Am.) 


metric  measurements  by  means  of  standard  data  from  color-matching  experiments,  so  that 
equal  color  sensations  can  be  specified  by  three  quantities  derived  from  the  objective 
stimulus. 

The  I.C.I.  (International  Commission  on  Hlumination)  system  is  that  most  used  for 

accurate  color  specification.  It  is 
based  upon  the  theory  that  the 
chromaticity  (hue  and  saturation) 
of  any  color  stimulus  can  be  speci- 
fied by  three  Quantities,  which  rep- 
resent the  proportions  of  three 
selected  primary  light  distributions 
that  are  necessary  to  match  the 
color  sensation  evoked  bj  a  given 
stimulus.  The  amounts  (luminosi- 
ties) of  each  of  three  selected  pri- 
mary color  sources  that  must  be 
added  together  in  order  to  match 
a  given  spectral  color  are  deter- 
mined by  these  color-matching 
experiments.  These  amounts  are 
called  the  tristimulus  vahies  of  the 
three  primaries  for  that  color. 
Three  standard  spectral  distribu- 
tions of  equal  energy,  whose  tri- 
stimulus  values  are  x  =  /*(X), 
$  »  /S(X),  and  s  =  /*(X)  (X  being 
wavelength),  are  especially  se- 
lected, with  dominant  wavelengths 
in  the  redT  green,  and  blue  respec- 
tively. Of.  Fig.  15.  These  pri- 
mary distributions  are  obtained  by 


400        440      480 


640      680       720 


520      560      600 
Wavelength,  rap 

FIG.  15.  Tristimulus  Values  for  the  Various  Spectral 
Colors.  The  values  x,  y,  z  are  the  amounts  of _  the  three 
I.C.I,  primaries  required  to  match  in  color  a  unit  amount 
of  energy  having  the  indicated  wavelength.  (Hardy,  H<md- 
book  of  Colarimetry.) 


transformations  from  the  experi- 
mental data  with  real  primaries; 
they  are  specially  selected  to  avoid 
negative  tristimulus  values  and  to 
meet  other  requirements  for  a  convenient  system.  The  y  spectral  distribution  function 
has  been  made  to  correspond  to  the  photopic  visibility  curve.  Any  homogeneous  spectral 
radiation  can  be  specified  by  the  tristimulus  values  read  directly  from  the  curves  or  special 
tables  (Hardy).  For  a  spectral  radiation  objectively  specified  by  a  spectrophotometric 


14-38 


OPTICS 


curve,  the  tristimulus  values  are  found  by  summing  up  the  products  of  the  energy  E\  for 
all  wavelengths  by  the  corresponding  tristimulus  values  from  the  standard  primaries  from 
the  tables,  viz., 


Through  the  choice  of  primary  standard  distributions  the  second  integral  gives  directly 
the  relative  luminosity  (brightness)  of  the  color  stimulus  on  a  black  (zero)  to  pure  white 
(100)  scale.  In  practice,  the  tristimulus  values  for  any  spectrophotometric  curve  are 
computed  by  averaging  the  products  of  xE\  (and  yE\  and  zJ?x)  for  equally  (spaced  wave- 
lengths. 

^  The  values  of  a;',  y',  and  zr  do  not  necessarily  measure  the  color  sensation,  but  they  do 
state  the  conditions  under  which  different  spectral  stimuli  will  result  in  the  same  color 
sensation.  It  must  be  borne  in  mind  that  this  sensation  may  vary  slightly  between  indi- 
viduals and  even  with  other  factors,  such  as  size  of  field  and  conditions  of  the  surround- 
ing fields. 

THE  CHROMATICITY  DIAGRAM.  A  convenient  graphical  representation  of  the 
chromaticity  of  any  spectral  stimulus  is  made  by  plotting  the  trichromatic  coefficients 
which  are  defined  by  the  ratios 

x'  ,  y' 

*  =  x>  +  y>  +  *>    ^    *"  *  +  y>  +  * 

and  the  third  would  be  related  to  the  other  two  by  x  +  y  +  z  —  1.  The  result  is  the 
standard  chromaticity  diagram  in  which  colors  of  equal  luminosity  are  represented  in 

terms  of  dominant  color  and 

°-9 ' '       saturation.  See  Fig.  16.  Color 

matches  are  known  to  be 
stable  through  wide  limits  of 
luminosity;  hencejthis  repre- 
sentation is  valid. 

The  visible  spectrum  ar- 
ranges itself  in  a  horseshoelike 
curve  beginning  with  red 
(700  mju)  and  passing  counter- 
clockwise through  orange, 
yellow,  green,  blue  to  a  deep 
purplish  blue  (400  m/z).  The 
straight  line  joining  the  ends 
of  the  spectrum  contains  the 
purples.  Clearly,  this  line  is 
not  part  of  the  spectrum. 
White  light  is  represented  by  " 
various  points  on  a  curve 
near  the  center  of  the  dia- 
gram, its  exact  position  de- 
pending upon  the  tempera- 
ture of  the  source.  The 
luminant  recommended  by 
the  International  Commission 
on  Illumination  is  shown  on 
the  diagram  at  C.  Any  given 
color  stimulus  will  also  be  rep- 
resented by  a  point  within 
this  curve.  Figure  17  shows 
the  approximate  color  names 


0.6 


0.7 


0.8 


FIG.  16.    The  Chromaticity  Diagram,  Showing  the  Geometrical 

Arrangement  of  the  Spectral  Colors  and  the  Locus  of  White  Lights 

(after  Hardy,  Handbook  of  Col&rimetry) 


for  the  different  portions  of  the  diagram.  The  dominant  wavelength  of  any  stimulus'will  be 
given  by  the  place  where  a  straight  line  drawn  through  C  and  the  point  intersects  the  nearest 
part  of  the  spectral  curve.  The  complementary  wavelength  will  be  the  intersection  point 
on  the  spectral  curve  diametrically  opposite.  The  saturation  or  excitation  purity  of  a 
given  color  can  be  determined  from  the  ratio  of  the  distances  of  the  point  from  C,  and 
from  the  dominant  wavelength  on  the  spectrum  curve.  The  resultant  color  of  a  mixture 
of  any  two  colors  represented  by  two  points  on  the  I.C.I,  diagram  will  be  represented 
somewhere  on  the  straight  line  drawn  between  the  two  points.  The  advantage  of  this 
representation  is  its  direct  quantitative  application  to  spectrophotometric  data.  Its 
disadvantage,  however,  lies  in  its  graphical  distortion  of  color  differences. 


THE  SPACE  SENSE 


14-39 


OTHER  SYSTEMS.  The  Ostwald  and  Munsell  systems  consist  of  orderly  arrange- 
ments of  colors  based  upon  their  visual  relationships,  and  not  dependent  upon  the  mixture 
of  pigments  or  upon  the  physical  measurements  of  the  light,  and  independent  of  the  illu- 
mination. Both  these  systems  are  illustrated  in  carefully  prepared  charts  with  orderly 
arrays  of  colored  patches,  which  vary  in  hue,  saturation,  and  value  according  to  the  con- 
cept of  the  author. 

COLOR  TOLERANCE.  Nearly  as  important  to  color  specification  is  the  necessity  of 
knowing  color  tolerances.  The  most  thorough  study,  to  date,  is  that  of  MacAdam  in 
which  the  color  tolerances  in 
changes  in  hue  and  in  saturation  °'9 
were  determined. 

COLOR  ADAPTATION.  If  the 
eyes  are  subjected  to  a  large  field 
of  a  spectral  color  for  a  period  of 
time,  the  perception  of  hue  of  other 
colors  becomes  modified  and  dis- 
torted. This  may  be  in  the  nature 
of  an  adaptation.  Two  electric 
lamps,  one  white  and  one  colored, 
say  red,  are  set  in  front  of  a  large 
white  screen  several  feet  apart. 
Then  an  object  is  placed  nearer 
the  screen  in  between  the  lamps, 
so  that  two  shadows,  one  from  the 
white  and  one  from  the  red  lamp, 
are  cast  upon  the  screen.  Upon 
continued  observation  it  will  be 
found  that  the  shadow  cast  from 
the  red  lamp  will  appear  green, 
complementary  to  the  red  color, 
while  the  shadow  from  the  white 
light  will  appear  red.  In  the  gen- 
eral perception  of  colored  objects 
in  fields  of  familiar  detail,  the  color 
relationships  tend  to  remain  the 
same,  in  spite  of  rather  wide  vari- 
ations of  illumination  intensities 
and  even  color.  If  the  visual  fields  are  isolated  and  confined,  however,  to  small  areas,  apart 
from  the  larger  field,  this  color-constancy  phenomenon  tends  to  disappear. 


0,1 


0.8 


FIG.  17.    Approximate  Color  Names  for  Various  Parts  of  the 
ChromaticityfDiagT&ra  (Kelley,  J.  Opticci  Soc~  Am.) 


12.  THE  SPACE  SENSE 

It  is  necessary  to  distinguish  clearly  between  objective  space,  filled  with  real  obje<yts, 
and  visual  space,  filled  with  visual  objects.  It  is  somewhat  meaningless  to  ask  whether  the 
position  of  visual  objects  is  identical  to  that  of  the  real  objects.  It  is  only  necessary  that 
the  relationships  between  visual  space  and  objective  space  be  ssifikaently  stable  so  that 
the  organism  can  effectively  operate  within  real  space.  Tbe  visual  perception  of  space  is 
essentially  egocentric  in  that  objects  are  localized  with  respect  to  the  body.  The  final 
perception  of  space  arises  through  a  complex  integration  of  (1)  the  visual  clues  inherent  in 
the  pattern  of  the  dioptric  images  that  fall  on  the  retinas,  (2)  the  simultaneous  impres- 
sions from  the  other  senses,  (3)  experiential  factors  that  have  been  associated  with  visual, 
auditory,  and  tactile  stimuli,  and  (4)  the  immediate  attention  of  the  individual.  It  is 
convenient  to  regard  space  perception  in  terms  of  (1)  the  discrimination  of  direetiom  (ego- 
centric and  bidimensional),  and  (2)  the  discrimination  of  depth,  that  is*  the  third  dimen- 
sion. 

DIRECTION  LOCALIZATION.  By  virtue  of  the  discrete  make-up  of  the  refea  (tfee 
retinal  mosaic),  the  various  parts  of  the  dioptric  images  on  the  retina  can  be  differentiated 
through  some  type  of  "local  signs"  associated  with  the  retinal  elements,  which,  in  this 
case,  is  a  subjective  visual  direction.  Resolving  power  and  visual  amity  are  measures  of 
the  keenness  with  which  this  differentiation  can  be  made.  The  fovea  is  the  primary 
point  of  reference,  and  the  subjective  direction  of  objects  is  described  in  terms  of  '^breadth" 
(right  or  left)  and  "height"  (above  or  below)  the  point  of  fixation  which  is  imaged  on  t&e 
fovea.  The  subjective  visual  directions  then  are  relative  to  the  fovea.  By  a  reSex  process 
associated  with  the  eye,  head,  and  body  movements,  the  change  in  the  subjective  direction 


14-40  OPTICS 

of  the  whole  visual  field  that  would  occur  with  eye  movements  is  counteracted,  so  that 
objects  tend  to  appear  in  the  same  "absolute'*  direction  in  spite  of  the  fact  that  the  images 
move  across  the  retina. 

Tbe  precision  of  the  relative  subjective  direction  varies  greatly  with  the  character  of 
the  visual  field,  and  in  many  cases  estimations  actually  are  inaccurate.  Probably  the 
highest  precision  is  in  the  estimation  of  when  lines  are  parallel  or  when  they  are  straight, 
and  in  the  comparison  of  angles  whose  corresponding  sides  are  parallel.  In  the  estimation 
of  differences  in  length,  a  greater  precision  is  found  in  the  horizontal  than  in  the  vertical. 
Vertical  distances,  moreover,  look  longer  than  equal  horizontal  distances.  In  the  estima- 
tion of  lengths,  accuracy  varies  with  the  lengths  to  be  compared  and  their  relative  posi- 
tions. When  attempting  to  determine  the  center  of  line  segments  without  eye  move- 
ments, the  eye  tends  to  overestimate  the  portion  on  its  own  side  (Kundt  partition  phe- 
nomenon), though  frequently  the  reverse  is  found.  Subdivided  and  nlled  spaces  look 
longer  than  unfilled  spaces.  Acute  angles  are  generally  overestimated  and  obtuse  angles 
are  underestimated.  Magnitudes  are  diminished  in  the  presence  of  larger  magnitudes 
and  magnified  in  the  presence  of  small  ones.  Many  of  the  well-known  illusions  are  exam- 
ples of  these  errors  in  the  perception  of  direction. 

VISUAL  ACUITY  AND  RESOLVING  POWER.  The  visual  acuity  of  the  eye  is  the 
degree  to  which  it  is  able  to  discriminate  fine  detail  in  the  visual  field,  and  this  varies 
considerably  with  the  type  of  detail,  the  contrast,  illumination,  surrounding  brightness, 
adaptation  level,  etc.  In  a  strict  sense,  resolving  power  is  more  definitive  than  visual 
acuity;  it  is  measured  as  the  reciprocal  of  the  smallest  visual  angle  (usually  in  minutes  of 
arc)  by  which  two  objects  can  be  seen  separately.  The  theoretical  limit  of  visual  resolution 
would  be  determined  by  the  size  of  the  diffraction  circles  on  the  retina  and  the  dimensions 
of  the  retinal  elements.  However,  for  pupils  of  normal  and  larger  sizes  the  spherical  and 
chromatic  aberrations,  together  with  small  irregular  astigmatism  that  is  usually  present, 
modify  the  nature  of  the  light  intensity  distribution  within  the  image,  so  that,  with  the 
contrasts  ordinarily  encountered,  the  resolution  is  better  than  that  based  upon  diffrac- 
tion alone.  For  small  pupils  a  marked  decrease  in  acuity  is  found,  probably  resulting 
from  the  increased  size  of  the  diffraction  circles,  but  for  pupils  larger  than  2  to  3  mm  the 
acuity  remains  nearly  constant  (Cobb).  Faint  stars  whose  angular  separation  is  1  to 
1  Vs  minutes  of  arc  can  usually  be  resolved.  With  small  point  light  sources  of  low  con- 
trast with  respect  to  the  background,  the  minimum  angle  of  resolution  may  be  100  seconds 
with  a  mean  error  of  5  seconds  of  arc.  With  repeating  patterns  such  as  lattices,  grids,  or 
checkerboard  patterns  the  minimum  angle  of  resolution  varies  between  50  and  75  seconds 
of  arc,  under  optimum  conditions. 

The  eye  is,  of  course,  capable  of  much  finer  discrimination  of  detail  than  that  which 
would  be  obtained  on  the  basis  of  resolving  power  alone.  Experiments  which  indicate 
this  fine  appreciation  of  detail  involve  least-perceptible  differences  in  contours.  These 
differences  have  been  expressed  as  the  mean  error  of  settings,  or  in  the  50/50  point,  in 
correct  judgments.  Typical  results  are  shown  in  the  table  below. 

Mean  Error, 
Test  seconds  Authority 

Widening  of  lower  half  of  slit 10-12  Wulfing  (1892) 

Coincidence  of  vertical  lines  (the  vernier) 8-12  Brian  and  Baker  (1912) 

Coincidence  of  vertical  lines  (the  vernier) 13  Best  (1900) 

Coincidence  of  vertical  lines  (the  vernier) . , . . , 3  Langland  (1929) 

Coincidence  of  vertical  lines  (the  vernier) 0.6  Trench  (1920) 

Coincidence  of  vertical  lines  (the  vernier) 2  Wright  (1942) 

Error  in  contact  of  white  disks  on  dark  background. .   15  Dale  (1920) 

Alignment  of  edges  of  two  rectangles 10  Hering  (1899) 

Error  of  setting  range  finders 2-6  von  Hof  e  (1920) 

Stereoscopic  displacement  of  images 5-7  Florian  (1930) 

In  general,  where  such  fine  discrimination  can  be  seen,  the  retinal  images  of  the  detail 
(such  as  extended  lines)  involve  the  activity  of  a  larger  number  of  retinal  elements.  Except 
for  monochromatic  yellow  light,  visual  acuity  is  decidedly  poorer  under  colored  illumi- 
nants  than  under  white  light.  This  is  especially  true  of  blue  and  only  slightly  less  so  under 
red  illuminations.  Part  of  this  latter  decrease  in  acuity  is  undoubtedly  related  to  the 
chromatic  aberration  of  the  eye. 

Visual  acuity  is  usually  measured  by  a  test  chart  of  letters  of  graded  sizes,  in  spite  of 
certain  obvious  inherent  faults.  Visual  acuity  is  considered  normal  when  letters  can  be 
identified  the  separation  of  adjacent  parts  of  which  subtend  an  angle  at  the  eye  of  1  minute 
of  arc.  A  line  of  letters,  each  of  which  subtends  a  5-minute  visual  angle,  is  printed  on  a 
chart  in  sequence  for  visual  distances  of  10,  15,  20,  25,  30,  40,  60,  100,  and  200  ft.  These 
letters  are  usually  arranged  with  the  smallest  at  the  bottom  of  the  chart,  and  with  a  single 


THE  SPACE  SENSE 


14-41 


large  E  at  the  top.  The  acuity  tests  are  usually  made  at  a  visual  distance  of  20  ft,  and 
that  line  of  letters  which  the  subject  can  just  read  is  then  recorded  relative  to  20  ft.  Thus 
20/20  represents  normal  vision;  20/15  indicates  that  print  which  could  ordinarily  be  read 
at  15  ft  can  be  read  at  20  ft  and  therefore  the  acuity  is  better  than  normal;  and  20/100 
indicates  that  that  type  which  should  be  legible  at  100  ft  can  be  read  only  at  20  ft  and 
therefore  the  acuity  is  much  less  than  normal.  Other  test  objects  frequently  used  are 
the  Snellen  hook,  the  Landolt  broken  ring,  and  checker-board  patterns. 

Because  of  the  increased  size  of  the  photosensitive  elements  toward  the  peripheral  parts 
of  the  retina  and  the  greater  number  of  them  associated  with  single  conductor  nerves, 
together  with  the  increased  magnitude  of  the  optical  aberrations  toward  the  periphery  of 
the  eye,  it  is  to  be  expected  that  the  resolving  power  of  the  eye  would  fall  off  rapidly 
toward  the  periphery.  Figure  IS  shows  this  decrease  in  resolving  power  with  the  increase 


70°    60°   50°   40°   30°   £0°    UD°     0°    10°   20°    30J   4O°    50°  60 

Temporal 


Fig.  IS.     The  Decrease  in  Visual  Acuity  toward  the  Peripheral  Parts  of  the  Retina  (Wertheim) 

of  the  peripheral  angle  (data  of  Wertheim).  A  very  rapid  decrease  in  acuity  occurs  to 
about  5  degrees,  and  from  there  the  decrease  is  much  slower.  This  loss  of  acuity  is  not  the 
same  in  all  the  meridians  of  the  eye  but  is  greater  in  the  vertical  meridian  than  in  the 
horizontal.  Figure  19  shows  the  isopters  of  the  retina,  that  is,  the  curves  of  equal  visual 
acuity  (data  of  Wertheim). 


90°f   80° 


70°  80C 


90* 


\ 


PIG.  19.    Isopters  of  Retina,  or  Curves  of  Equal  Visual  Acuity,  Showing  that  the  Decrease  in  Acuity  is 
Greater  in  the  Vertical  Meridian  than  in  the  Horizontal  (Wertheim) 


14-42 


OPTICS 


There  has  been  evidence  that  visual  acuity  is  greater  for  distant  vision  than  for  near 
vision.  This  phenomenon  is  believed  to  be  dependent  upon  the  type  of  test  and  the  condi- 
tions under  which  the  test  is  made,  especially  for  peripheral  vision,  because  opposite  results 
have  been  reported. 

IRRADIATION.  Owing  to  the  aberrations  of  the  optical  system  of  the  eye,  the  images 
of  discrete  points  are  not  denned  with  sharp  boundaries.  Rather,  the  intensity  of  the 
light  in  the  image  falls  off  in  a  bell-shaped  curve.  The  perceived  contour  depends  upon 
the  differential  sensitivity  of  the  eye,  but  the  image  always  corresponds  to  an  object  larger 
than  the  original.  The  position  of  the  perceived  contour  will  tend  to  be  toward  the  less 
intense  end  of  the  light  distribution  curve.  Accordingly,  bright  objects  seen  against  a 
dark  background  appear  larger  than  dark  objects  of  equal  size  against  a  bright  background. 
This  phenomenon,  called  irradiation,  depends  upon  the  relative  luminosity  of  the  adjacent 
surfaces  and  varies  with  individuals  because  of  differences  in  ocular  aberrations.  Objects 
of  large  angular  size  are  increased  proportionately  only  to  a  small  extent,  and  their  size 
can  be  said  to  remain  constant.  For  small  objects,  however,  the  irradiation  increases  with 
decrease  in  angular  size.  On  the  other  hand,  the  separation  of  small  black  lines  against 
a  bright  background  may  actually  appear  larger  than  it  is.  This  so-called  negative  "irra- 
diation" is  explained,  however,  by  the  influence  of  the  mechanism  of  simultaneous  contrast, 
whereby  the  apparent  brightness  (or  color)  of  an  area  is  influenced  (enhanced)  by  adjacent 
areas  of  different  brightness  (or  color) . 

VISUAL  ACUITY  AND  ILLUMINATION.  The  visual  acuity  of  the  fovea  of  the  eye 
increases  markedly  with  an  increase  in  the  luminosity  of  the  field,  but  that  of  the  periph- 
eral retina  changes  scarcely  at  all.  In 
general,  then,  studies  on  visual  acuity 
and  illumination  pertain  to  foveal 
vision.  The  data  of  Konig  illustrated 
in  Fig.  20  (as  recomputed  by  Hecht) 
are  among  the  most  complete  on  this 
phenomenon.  They  were  obtained 
with  a  black  test  object  upon  a  white 
background  and  are  therefore  of  maxi- 
mum contrast.  For  illuminations 
from  0.013  to  10  lumens  per  sq  ft  the 
visual  acuity  is  approximately  propor- 
tional to  the  logarithm  of  the  lumi- 
nosity. The  visual  acuity  obtained  at 
lower  illuminations  decreases  rapidly 
in  the  neighborhood  of  0.001  lumen 


1.8 


1.6 


1.4 


1*2 
fl.O 

4? 

§0.8 


0.6 


0.4 


0.2 


-5       -4      -3 


2-1          0          1 
Log  I-mnniamberts 


per  sq  ft,  where  scotopic  vision  begins. 
From  here,  for  lower  illuminations, 
only  rod  vision  is  involved  and  the 
change  in  visual  acuity  is  small.  Al- 
though Konig's  data  show  a  maximum 
for  illuminations  greater  than  100 

FIG.  20.    The  Relation  between  Visual  Acuity  of  the   Iumens  Per  sq  ft,  and  thereafter  a  con- 
Fovea   and   Illumination   (Data  of   Koenig  revised  by    stant  visual  acuity,  more  recent  data 
Hecht,  J".  Gen.  Physiol.)  have  indicated  that  this  maximum  is 

,,     -  ,,  ,     ,      ,  ,  due  to  the  fact  that  the  brightness  of 

tne  neld  surrounding  the  background  of  the  test  object  was  low  compared  to  that  of  the  test 
object  itself.  In  experiments  where  the  brightness  of  the  surrounding  field  is  the  same  as 
that  of  the  background  for  the  test  object,  a  maximum  is  not  found  and  the  visual  acuity 
continues  to  increase  at  only  a  slightly  lower  rate  with  a  continued  increase  of  illumina- 
tion. The  visual  acuity  is  lower  for  colored  light  than  for  white  of  the  same  brightness, 
being  poorer  in  the  order  green,  red,  blue.  In  general,  visual  acuity  is  also  reduced  under 
conditions  of  glare. 

When  detail  is  exposed  for  short  intervals  of  time  (less  than  about  1  sec),  and  the 
luminosity  of  the  background  remains  constant,  the  visual  acuity  varies  roughly  with 
the  logarithm  of  the  time  of  exposure.  Below  exposure  times  of  about  0.01  to  0.03  sec, 
depending  upon  the  illumination,  the  product  of  time  and  illumination  is  constant  for  the 
same  visual  acuity. 

In  Konig>  data  the  contrast,  defined  as  the  ratio  of  the  difference  in  luminosity  of  the 
background  and  the  test  object  to  the  luminosity  of  the  background,  was  very  high.  It 
has  been  shown  that  with  lower  contrasts  the  visual  acuity  is  also  lower.  Figure  21  illus- 
trate data  from  Cobb  and  Moss  on  the  relationship  between  contrast  and  visual  acuity. 
-No  data  are  available  for  bright  detail  against  a  darker  background. 


THE  SPACE  SENSE 


14-43 


MINIMUM  VISIBLE.  Similar  to  the  problem  of  visual  resolution  is  that  of  the  mini- 
mum visible.  What  is  the  angular  size  of  the  smallest  detail  that  can  be  perceived?  As 
in  resolving  power,  this  depends  upon  an  intensity 
discrimination  within  the  image  on  the  retina  and 
hence  varies  with  the  nature  of  the  dioptric  image, 
the  contrast  sensitivity  of  the  eye,  the  size  and 
form  of  the  detail,  exposure  time,  etc.  Lines  are 
more  readily  seen  than  dots,  and  repeating  patterns 
of  dots,  lines,  etc.,  are  perceived  even  more  readily. 
Small  voluntary  and  involuntary  eye  movements 
also  aid  in  the  discrimination  of  fine  detail.  Under 
general  illumination  a  line  against  a  bright  back- 
ground can  be  seen  when  its  width  subtends  a  visual 
angle  of  3  to  4  seconds  of  arc.  Hecht  and  Mints 
found  that  with  a  similar  test  object  the  angular 
width  of  a  wire  that  could  just  be  seen  varied  from 
10  minutes  to  0.5  second  over  the  complete  range 
of  illuminations.  The  rninimum  value  0.5  second 
represents  a  discrimination  of  about  1  to  2  per  cent 
difference  in  light  intensity  in  the  image  on  the 
retina.  When  a  narrow  illuminated  slit  that  is  ex- 
posed for  short  durations  is  just  visible,  it  is  found 
that  (visual  angle)  X  (duration  of  exposure)  X 
(adapting  luminosity  of  background)  is  approxi- 
mately a  constant  for  flashes  of  0.004  to  0.189  sec 
of  duration  (Niven  and  Brown). 

As  indicated  above  it  is  now  believed  that  the 
rate   at    which    contrast   sensitivity  and   visual 


100 


21.     Relationship  between  Minimum 
Angle  of  Resolution,  Luminosity  of  Back- 
ground, and  the  Contrast  of  the  Test  Ob- 
ject Relative  to  its  Background  (Cobb  and 
Moss,  J.  Frankiin  Im&itvle) 


acuity  in  the  absence  of  glare  increase  with  luminosity  of  background  is  not  appreciably 
diminished  with  the  very  high  luminosities  (cf.  Figs.  8  and  20).  In  other  words,  there 
is  no  optimum  luminosity.  The  question  arises  then  as  to  what  the  general  illumination 
should  be  for  adequate  vision.  Although  this  question  cannot  be  answered  specifically 


B 

Relative  Visual  Prof  IcJancy-  Par  cent 
to.bg)  Co  to  to  to  <p  q 

^O  OOO  O  0  4*  CO  do  3 

/ 

/ 

/ 

Contrast  Sen 

sitlvlU  / 

7        x 

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X' 

Minimum  VI 

1W9^ 

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al  Acuity 

/ 

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iteriorsx 

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&•*              a                10               io2             ios              10*              io6              ic 

Luminosity  of  Backgrocrad-Lamens/Sq.Ffc. 
Fia.  22.    Relative  Visual  Proficiency  for  Contrast  Sensitivity,  the  Minimum  Visible,  and  Visual  Acuity, 
as  a  Function  of  Brightness  of  Background  (after  Moon  and  Spencer,  J,  Optical  See.  Am.) 

it  is  of  value  to  know  the  relative  degree  of  visual  efficiency  for  any  given  general 
illumination.  A  scheme  for  indicating  this  has  been  suggested  by  Moon  and  Spencer.  In 
contrast  sensitivity,  the  quantity  Tc  is  denned  as  the  ratio  of  the  contrast  sensitivity 


14-44 


OPTICS 


(J/AI)  for  the  given  illumination  (I)  to  the  maximum  contrast  sensitivity  that  would  be 
obtained  with  uniform  fields  and  very  high  daylight  luminosities.  It  represents  the  rela- 
tive proficiency  of  contrast  discrimination  at  any  given  illumination.  Similar  ratios  Tv  and 
Tc  are  defined  for  the  minimum  visible  and  for  visual  acuity.  In  Fig.  22  the  theoretical 
values  of  each  of  these  three  visual  functions  are  shown  as  dependent  upon  the  luminosity 
of  the  background.  For  outdoor  lighting  the  relative  visual  proficiency  would  range 
between  70  and  95  per  cent,  and  for  interiors  with  artificial  lighting  it  would  probably  be 
less  than  80  per  cent. 

PERCEPTION  OF  MOTION.  The  motion  of  objects  in  space  is  perceived  as  motion 
relative  to  the  observer  or  as  relative  motion  between  the  objects  themselves.  Under 
special  circumstances  the  motion  of  objects  seen  with  respect  to  each  other  and  to  the 
observer  may  be  equivocal,  for  example,  the  apparent  motion  of  the  moon  observed  through 
moving  clouds.  In  a  field  of  large  and  small  objects  the  larger  objects  are  said  to  exhibit 
more  stability  and  are  less  liab.le  to  apparent  motion,  while  small  objects  exhibit  apparent 
movement  more  readily  and  are  said  to  be  more  mobile.  Moreover,  objects  imaged  on  the 
peripheral  parts  of  the  retina  appear  less  mobile  than  those  falling  on  the  fovea. 

Perception  of  motion  rests  upon  a  temporal  reaction  to  stimuli  falling  successively  on 
neighboring  points  of  the  retina,  but  its  appreciation  appears  only  partially  related  to 
resolving  power.  It  is  thought  to  be  a  visual  sensation  resulting  from  experience.  For 
objects  whose  visual  angle  of  movement  is  small,  motion  is  inferred  from  the  apparent 
positions  after  lapses  of  time,  for  example,  a  moving  train  seen  at  a  great  distance.  The 
true  perception  of  motion,  however,  is  a  specific  sensation,  with  both  lower  and  upper 
thresholds  of  discrimination.  The  impression  of  motion  can  also  occur  (within  definitely 
prescribed  limits)  from  successive  stimuli  arising  from  separated  stationary  sources.  This 
is  called  apparent  movement  by  psychologists. 

The  threshold  of  the  perception  of  true  motion  relates  to  the  least  angular  ^displacement 
of  an  object  that  can  be  recognized  in  unit  time.  With  comparison  objects  in  the  field  of 
view  the  lower  threshold  is  1-2  minutes  of  arc  per  second.  Without  comparison  objects 
this  value  must  increase  10  to  20  times.  The  least  angular  movement  that  can  be  detected 
between  two  fixed  points  is  sometimes  called  the  movement  acuity;  under  ordinary  cir- 
cumstances, this  is  found  to  be  10  to  20  seconds  of  arc,  when  stationary  comparison 
objects  are  in  the  field  of  view.  Without  comparison  objects  this  increases  to  more  than 

1  minute  of  arc.  When  the  range  of 


70 


|  60 

*0 

JJ50 


A40 
I 

I2 

5 
J20 

llO 


Form  Acuity- 


Blind 
Spot 


Motion  Acuity 


the  movement  (the  distance  between 
the  beginning  and  end  points)  is  fixed, 
the  shortest  lapse  of  time  in  which 
motion  is  seen  has  also  been  measured. 
Under  ordinary  circumstances,  for  ex- 
ample, the  threshold  for  a  10~degree 
range  varies  between  0.027  and  0.079 
sec.  The  upper  limit  for  the  discern- 
ment of  motion,  when  the  sensation 
becomes  a  meaningless  blur,  is  1.4  to 
3.5  degrees  per  Vioo  sec.  The  move- 
ment acuity  decreases  with  increase  in 
illumination. 

The  threshold  for  the  discrimination 
of  movement  is  least  at  the  fovea  and 
increases  rapidly  out  toward  the  periph- 
ery of  the  retina,  though  it  has  been 
thought  that  the  increase  is  much  less 
than  the  decrease  of  visual  acuity  to- 
ward the  periphery.  Figure  23  illus- 
trates the  data  of  Klein,  which  show, 
however,  that  the  acuity  and  motion 
thresholds  are  essentially  the  same  ex- 
cept at  the  fovea.  It  appears  that  the  discrimination  of  motion  is  greatest  at  the  fovea, 
but  the  appreciation  of  motion  is  most  marked  in  the  peripheral  retina,  to  a  degree  be- 
yond the  actual  resolving  power.  The  discrimination  of  movement  in  the  periphery  jis 
somewhat  poorer  above  and  below  the  fovea  than  to  the  right  or  left.  There  is  a  tendency 
to  overestimate  the  extent  of  motion  in  the  periphery,  and,  furthermore,  the  subjective 
direction  of  that  motion  is  somewhat  vague  and  uncertain.  However,  the  sense  of 
movement  is  highly  developed. 

THE  PHI-PHENOMENON.     The  impression  or  illusion  of  movement  can  occur  in 
certain  situations  without  there  being  actual  physical  movement.    The  most  important  of 


20          30          40          50 
Peripheral  Angle- Degrees 


70 


FIG.  23.  The  Thresholds  for  Form  Acuity  and  Motion 
Acuity  for  Same  Test  Objects.  The  motion  acuity  is 
given  in  terms  of  the  minimum  distance  required  for 
perception  of  motion.  (Data  of  Klein,  Arch.  Psychol.) 


THE  SPACE  SENSE  14-45 

the  apparent  movements  is  known  in  its  purest  form  as  the  p&i-phenomeiaa,  whereby 
apparent  movement  can  occur,  under  certain  conditions,  when  separated  stationary  points 
are  illuminated  successively.  Usually  the  illusion  of  movement  can  be  distinguished  from 
real  motion,  however,  and,  as  such,  may  be  variously  interpreted  by  different  observers. 
Only  in  the  neighborhood  of  an  angular  succession  of  20  degrees  per  second  are  tlie  two 
motions  difficult  to  identify  (de  Silva).  For  the  impression  of  motion  there  is  a  time  rela- 
tionship between  the  duration  of  the  first  stimuli,  the  separation  of  stimuli,  the  interval 
between  successive  stimuli,  and  the  brightness  of  background,  the  impression  depending 
also  to  a  great  extent  upon  the  attitude  of  the  observer.  Under  ordinary  conditions,  with 
a  dark  background,  it  has  been  found  that,  for  two  point  stimuli  1  cm  apart,  the  duration 
of  each  of  which  is  0.05  sec:  (1)  if  the  two  stimuli  are  nearly  instantaneous  (0.03  sec)  they 
will  appear  simultaneous;  (2)  if  the  time  interval  is  greater  than  0.2  sec  the  points  will 
appear  discrete  and  stationary;  (3)  for  time  intervals  intermediate  between  these  two 
there  will  be  an  apparent  movement  of  the  first  point  to  the  second,  with  an  optimum  at 
0.06  sec.  Of  two  light  points  of  different  intensity  the  weaker  tends  to  move  toward  the 
brighter.  Other  examples  of  apparent  movement  are  in  the  cinema,  in  neon  signs,  and  in 
stroboscopic  instruments. 

THE  PERCEPTION  OF  DEPTH.  In  the  perception  of  depth  one  must  distinguish 
between  the  perception  of  depth  differences  (how  much  one  object  appears  in  front  of  or 
behind  another)  and  the  absolute  localization  by  which  the  actual  distance  of  objects 
from  the  individual  is  estimated.  Binocular  vision,  through  the  phenomenon  of  stereopsis, 
provides  the  most  accurate  means  of  relative  depth  discrimination. 

The  absolute  localization  of  objects  in  space  results  from  a  complex  process  that  involves 
both  monocular  and  binocular  perception.  It  has  been  usual  to  state  that  the  perception 
of  depth  by  monocular  vision  is  a  conception  of  depth  (distance)  attained  through  expe- 
rience with  certain  relationships  (clues)  that  will  exist  between  parts  of  the  retinal  images 
of  different  objects  in  space.  The  more  important  of  these  visual  clues  are:  (a)  Overlay, 
by  which  the  images  of  near  objects  overlap  and  tend  to  hide  those  of  the  naore  distant 
objects,  (b)  Perspective,  which  depends  upon  the  fact  that  objects  of  eo^ial  size  have 
smaller  retinal  images  when  at  a  distance  than  when  near  by.  linear  perspective  relates 
to  the  apparent  convergence  of  parallel  lines  that  recede  in  the  distance  (railroad  tracks, 
etc.).  Details  within  known  objects  are  more  readily  seen  when  near  than  when  distant. 
Thus,  the  size  of  retinal  image  related  to  known  size  provides  the  clue  for  estimation  of 
distance,  (c)  Aerial  perspective,  through  which  the  edges  of  objects  at  a  distance  are  leas 
clearly  denned  than  those  near  by.  Moreover,  the  more  distant  objects  appear  cooler 
(bluer)  in  color  on  account  of  atmospheric  haze.  Near  objects  appear  brighter  with  more 
color  saturation  than  those  more  distant,  (d)  Light  arid  $hadou\  which  give  clues  as  to 
shapes  and  relative  positions  of  objects,  (e)  Parallax,  which  results  from  head  movements, 
for  the  relative  alignment  of  more  distant  objects  changes  less  than  that  for  near  objects 
with  the  same  movement.  This  clue  to  depth  perception  is  very  strong,  and  the  precision 
of  depth  estimation  through  it  is  nearly  as  great  as  that  of  stereoscopic  depth  perception 
(Tschermak) .  (/)  Height,  whereby  objects  seen  above  others  are  also  judged  more  distant. 
(g)  To  a  small  extent  accommodation  and  convergence,  through  a  proprioceptive  sens* 
arising  from  the  muscles  of  the  eyes,  may  provide  a  due  for  gross  differences  in  depth. 

As  is  well  known  it  is  impossible  to  present  a  single  bidimensional  picture  which  will 
have  all  the  characteristics  of  the  actual  three-dimensional  scene  being  portrayed.  This 
is  especially  true  when  the  picture  is  being  viewed  binocularly.  As  the  eyes  move  over  a 
painting,  no  change  in  accommodation  or  convergence  is  demanded  and  obviously  no  dis- 
parity clues  are  present  for  stereoscopic  space  localisation  of  the  details  in  the  picture  as 
there  would  be  in  the  actual  scene.  The  perspective  in  a  pictorial  representation  of  any 
scene,  whether  a  photograph  or  a  painting  (except  in  certain  art  styles) ,  refers  to  a  fixed 
station  point  and  in  order  to  appear  correct  must  be  viewed  from  that  point.  A  contact- 
print  photograph  should,  then,  be  viewed  at  that  distance  that  approximates  the  focal 
length  of  the  camera  that  took  the  photograph.  Viewed  at  other  distances  the  perspective 
is  exaggerated  and  unnatural.  The  viewing  distance  of  an  enlarged  photograph  will  be 
equal  to  the  product  of  the  focal  length  of  the  camera  and  the  magnification  of  the  en- 
largement. This  larger  viewing  distance  explains  the  more  pleasing  effect  derived  from 
looking  at  enlarged  photographs.  It  is  desirable  that  pictures  be  viewed  in  such  a  way 
that  the  observer  feels  himself  a  part  of  the  scene.  The  screen  or  plane  of  the  picture 
itself  should  seem  detached  from  the  surroundings  and  preferably  should  appear  indef- 
initely localized.  To  accomplish  this  is  not  always  easy;  usually  it  can  be  only  approx- 
imated, especially  for  pictures  viewed  near  by  with  binocular  vision.  It  must  be  pointed 
out,  however,  that  with  the  proper  attitude  on  the  part  of  the  observer,  and  in  the  absence 
of  distracting  peripheral  detail,  the  several  psychological  constancy  phenomena  tend  to 
correct  small  distortions  of  form,  of  size,  and  of  the  color-brightness  relationships. 


14-46 


OPTICS 


Right  Eye 


13.  BINOCULAR  VISION 

Binocular  vision  is  the  coordinated  use  of  the  two  eyes,  in  which  a  single  perception  of 
external  space  is  obtained,  and  by  which  the  specific  sensation  of  stereoscopic  depth  per- 
ception is  made  possible.  The  final 
perceptual  images  from  the  two  eyes 
are  normally  said  to  fuse  in  the  brain, 
and  through  the  strength  of  the  fusion 
impulse  all  movements  of  the  eyes 
become  coordinated  in  the  process  of 
fixating  different  points  in  space. 

One  differentiates  between  the  bin- 
ocular visual  field,  which  will  cor- 
respond to  that  portion  of  visual  space 
for  which  the  images  of  the  two  eyes 
will  overlap,  and  the  binocular  field  of 
fixation  which  is  the  maximum  field 
swept  over  by  the  movements  of  the 
two  eyes.  Figures  24  and  25  illustrate 
the  approximate  angular  dimensions 
of  these  fields. 

The  luminosity  of  objects  is  some- 
what increased  by  the  use  of  two  eyes 
over  that  of  only  one,  depending  upon 

a  ^  ^  experimental  conditions.  Also  the 

100  visual  acuity  of  two  eyes  is  usually 

PIG.  24.    The  Binocular  Visual  Field  (after  Southall, 
Introduction  to  Physiological  Optics) 


Left  Eyi 
Field 


found  to  be  higher  than  that  of  one  eye. 
FUSIONAL  AREAS.  When  the 

eyes  are  fixated  upon  one  point  in 

space,  not  all  other  points  in  space  are  perceived  single.  In  fact,  only  those  points  will  be 
seen  single  that  are  situated  within  a  certain  three-dimensional  region  determined  by  the 
distance  of  the  fixation  point  and  by 

Region  of  Single 
Binocular  Vision 


the  equivalent  anatomical  extent  of 
certain  areas  on  the  retinas  (Panum's 
fusional  areas).  See  Fig.  25.  Points 
in  space  outside  these  areas  would 
ordinarily  be  seen  double  (physiologi- 
cal diplopia);  however,  unless  atten- 
tion is  called  to  them,  one  is  usually 
suppressed.  The  functional  extent  of 
Panum's  area  varies  to  some  extent 
with  visual  conditions,  and  its  meas- 
urement of  size  decreases  somewhat 
with  practice.  The  size  increases 
away  from  the  foveas  toward  the  pe- 
riphery. Near  the  fovea  its  minimum 
extent  is  about  6  to  12  minutes  of  arc 
in  the  horizontal  meridian  and  about  6 
minutes  of  arc  in  the  vertical  meridian. 
All  objects  beyond  a  fixation  point  at 
about  50  ft  from  the  observer  will  gen- 
erally be  seen  single.  Double  images 
exert  compulsion  in  nervations  for  the 
eyes  to  move  so  as  to  overcome  the 
doubling.  In  the  horizontal  meridian, 
the  actual  eye  movements,  however, 
are  usually  subject  to  the  will. 

STEREOSCOPIC  VISION.  Stere- 
oscopic vision  rests  upon  the  fact  that 
each  of  the  two  eyes,  by  virtue  of 
their  separation,*  sees  objects  in  the 
visual  field  from  a  slightly  different 


Inferred  Cortical  -^ 
Coansctions--^/ 


FIG.  25.    The  Spatial  Region  of  Single  Binocular  Vision 
and  the  Geometrical  Relations  for  the  Disparity  of  Reti- 
nal Images  in  the  Two  Eyes 


*  The  variation  of  the  interpupillary  distance,  in  the  general  population,  is  between  55  and  75  mm, 
•with  a  median  at  about  63  mm  (measured  while  the  eyes  are  looking  at  a  distant  object). 


BINOCULAB  VISION 


14-47 


point  of  view,  and  hence  the  two  retinal  image  patterns  arc  slightly  dissimilar.  See 
Fig.  26.  The  angular  separation  of  the  retinal  images,  arising  from  two  points  ia 
space,  will  be  different  in  the  two  eyes,  if  one  of  the  points  is  farther  away  from  the  ob- 
server. This  difference  in  angular  separation  defines  a  disparity  between  the  images.  A 
disparity  must  always  relate  to  two  points  in  space.  Referring  again  to  Fig.  25,  the  double 
images  from  points  at  A  and  beyond  are  said  to  be  uncrossed  disparate  with  respect  to  the 
images  of  F,  for  if  the  right  eye  is  suddenly  closed  the  right  half-image  vanishes.  Similarly 
the  images  of  points  at  B  and  nearer  are  said  to  be  crossed  disparate  with  respect  xo  the 
images  of  F,  for  if  the  right  eye  is  suddenly  closed  the  left  half-image  vanishes-  Between 
the  points  A  and  B  there  will  be  some  position  for  which  the  images  from  the  two  eyes 
will  not  be  disparate,  either  crossed  or  uncrossed  with  respect  to  the  Image  of  F.  Tins 
criterion  defines  the  position  of  points  on  the  horopter  surface.  Stereoscopic  depth  per- 
ception arises  by  virtue  of  the  disparity  between  retinal  images,  uncrossed  disparity  being 
associated  with  the  sense  of  distance  away  from  the  fixated  point.  Objects  seen  in  crossed 
disparity  are  seen  nearer  than  that  point.  Stereopsis  is  a  specific  sensation  resting  upon 


,  Pyramid  In  Space 


R.E, 


FIG.  26,     Illustration  of  the  Difference  in  the  Image  Patterns  of  the  Two  Eyes  in  Binocular  Space 

Perception 

the  physiological  and  anatomical  organization  of  the  retinal  elements  of  the  two  eyes.  It 
usually  occurs  immediately  for  almost  instantaneous  illumination,  and  it  exists  within 
the  entire  binocular  visual  field. 

The  angular  disparity  between  the  images  of  two  separated  objects  in  depth  in  space 
will  increase  with  the  separation  of  the  eyes  and  will  decrease  with  their  distance  from 
the  eyes.  Though  a  sensation  of  greater  depth  will  be  associated  with  greater  disparities 
of  the  images,  the  effect  is  not  rigidly  geometrical.  For  a  given  angular  disparity,  the 
apparent  depth  interval,  to  be  quantitative,  must  vary  with  the  square  of  the  vis- 
ual distance. 

The  visual  acuity  of  the  poorer  eye  must  ultimately  limit  the  threshold  of  stereoscopic 
vision.  Likewise,  the  threshold  of  stereoscopic  vision  will  depend  upon  those  factors 
influencing  visual  acuity  and,  like  visual  acuity,  will  vary  with  the  duration  of  the  stimuli. 
For  central  vision  and  for  durations  longer  than  3  sec,  maximum  acuity  is  obtained;  for 
shorter  durations  down  to  0.2  sec  there  is  a  4  to  5  fold  increase  hi  threshold,  after  which 
the  instantaneous  threshold  is  nearly  constant  (Langlands). 

Although  varying  with  interpupillary  distance  and  with  individuals  as  well  as  with  the 
nature  of  the  detail  in  the  visual  field,  the  limiting  threshold  for  stereopois  frequently  is 
taken,  on  the  average,  as  30  seconds  of  arc.  There  is,  correspondingly,  a  visual  distance 
beyond  which  objects  at  greater  distances  cannot  be  seen  stereosoopicilly.  This  limiting 
distance  will  roughly  be  450  meters,  or  about  a  quarter  of  a  mile.  Under  certain  condi- 
tions, when  the  threshold  of  stereopsis  may  be  as  low  as  6  seconds  of  arc,  this  distance  will 
be  exceeded  several  times*  For  the  peripheral  regions  of  tlie  retina,  this  limiting  distance 
will  be  much  reduced,  owing  to  the  lowered  aexiity. 

Two  points  separated  vertically  will  have  images  in  the  two  eyes  which  are  vertically 
disparate  if  those  points  are  located  to  the  right  or  left  of  the  plane  perpendicular  to  the 
interpupillary  base  line.  Vertical  disparities  do  not  give  rise  to  the  perception  of  depth, 
as  do  horizontal  disparities,  but  they  undoubtedly  aid  in  the  spatial  localization  of  objects. 


14-48  OPTICS 

Since  the  invention  of  the  stereoscope  by  Wheatstone  in  1838,  and  the  discovery  of  the 
bases  for  stereoscopic  vision,  various  instruments  have  been  devised  whereby  stereoscopic 
views  could  be  obtained  with  pictures.  In  general,  the  procedure  consists  in  presenting 
slightly  different  pictures  or  drawings  before  the  two  eyes  which  would  correspond  to  the 
views  that  each  of  the  eyes  would  perceive  if  they  had  been  present  when  the  photograph 
was  taken.  The  devices  include  the  mirror-  and  prism-stereoscopes,  the  haploscope,  the 
red  and  green  and  polaroid  anaglyphs,  grid  devices,  and  even  motion  pictures  where  the 
left-  and  right-eye  views  are  projected  alternately.  The  problem  of  projecting  pictures 
for  stereoscopic  vision,  so  as  to  preserve  correct  disparity  relationships  and  the  correct 
perspective  features,  is  a  difficult  one,  especially  for  a  group  of  observers  (Rule) . 

RIVALRY.  If  the  fields  presented  to  the  two  eyes  are  greatly  different,  for  example 
in  radically  different  colors  or  in  detail,  instead  of  fusion  or  even  a  simultaneous  percep- 
tion of  both  patterns  taking  place,  retinal  rivalry  occurs.  In  this,  either  one  field  or  the 
other  is  seen,  usually  alternately.  In  the  case  of  dissimilar  patterns  sometimes  sections  of 
each  are  seen  simultaneously,  but  seldom  both  in  the  same  region  of  the  visual  field.  The 
period  of  alternation  of  the  two  visual  fields  varies  between  2  and  12  sec,  depending  upon 
differences  in  luminosity,  area,  distinctness  of  detail  within  the  fields,  and  central  or 
peripheral  vision.  One  field  may  prevail  over  the  other  for  longer  periods  if  there  is  a 
great  difference  in  luminosity,  or  if  intelligible  detail  exists  on  one  and  not  the  other,  etc. 
Ocular  dominance  may  also  be  a  factor  in  the  field  that  prevails  the  longer.  Only  in  the 
case  of  certain  color  differences  can  fusion  and  therefore  the  emergence  of  a  mixed  color 
arise. 

BIBLIOGRAPHY 

General 

Eelmholtz,  H.  v.,  Physiological  Optics,  Eng.  trans,  by  J.  P.  C.  Southall.    Optical  Soc.  Am.  (1925). 
Duke-Elder,  W.  S.,  Text-Book  of  Ophthalmology,  Vol.  1.     Kimpton,  London  (1932). 
Polyak,  S.  L.     The  Retina.     University  of  Chicago  Press  (1941). 

Walls,  G.,  The  Vertebrate  Eye.     Cranbrook  Institute  of  Science,  Bull.  19  (August  1942). 
Southall,  J.  P.  C.t  Introduction  to  Physiological  Optics.     Oxford  University  Press  (1937). 
Emsley,  H.  H.t  Visual  Optics,  3d  Ed.    Hatton  Press,  London  (1944). 
Bartley,  S.  H.t  Vision,  A  Study  of  Its  Basis.     Van  Nostrand  (1941). 
Guilford,  J.  P.,  Psychometric  Methods,  Chapters  IV-VI.     McGraw-Hill  (1936). 
Luckiesh,  M.,  and  F.  K.  Moss,  The  Science  of  Seeing.     Van  Nostrand  (1937). 
Hardy,  A.  C.,  Handbook  of  Cplorimetry.     Mass.  Inst.  of  Technology  (1936). 
Moon,  P.,  The  Scientific  Basis  of  Illuminating  Engineering,  Chapter  XII.    'McGraw-Hill  (1936). 
Woodworth,  R.  S.,  Experimental  Psychology,  Chapters  22-26.     Henry  Holt  (1938). 
Herirfg,  E.,  Raumsinn,  Hermans  Handbuch  d.  Physiol.  d.  Sinnesorgane.     An  English  translation,  Spatial 
Sense  and  Movements  of  the  Eye,  C.  A.  Radde.     American  Academy  of  Optometry,  Baltimore  (1942). 

REFERENCES 

Asher,  L.,  Monoculares  und  binoculares  Blickfeld  eines  Emmetropen,  Arch.  f.  Ophth.,  Vol.  48,  427- 

431  (1899). 
Bond,  M.  E.t  and  D.  Nickerson,  Color-order  Systems.    Munsell  and  Qstwald,  J.  Optical  Soc.  Am., 

Vol.  32,  709  (December  1942). 
Cobb,  P.  W.,  The  Influence  of  Pupillary  Diameter  on  Visual  Acuity.     Am.  J.  PhysioL,  Vol.  36,  335 

(1915). 
Cobb,  P.  W.,  and  F.  K.  Moss,  The  Four  Variables  of  the  Visual  Threshold,  J.  Franklin  Inst.,  Vol.  205, 

831  (June  1928). 
Cobb,  P.  W.,  and  F.  K.  Moss,  Glare  and  the  Four  Fundamental  Factors  of  Vision,  I.E.S.  Trans.,  Vol. 

23,  1104  (1928). 
De  Silva,  H.  R..  An  Experimental  Investigation  of  the  Determinants  of  Apparent  Visual  Movement, 

Am.  J.  Psych.,  VoL  37,  461-501  (1926). 
Dohner,  D.  R.,  and  C.  E.  Foss,  Color-mixing  Systems.     Color  vs.  Colorant  Mixture,  J.  Optical  Soc. 

Am.,  Vol.  32,  702  (December  1942). 
Duane,  A.,  Studies  in.  Monocular  and  Binocular  Accommodation,  etc.,  Am.  J.  OphtJial.,  3rd  Series, 

Vol.  5,  865  (1922). 

Engstrom,  E.  W.,  A  Study  of  Television  Image  Characteristics,  Pt.  II,  Proc.  I.R.E.,  Vol.  23,  295  (1935). 
Freeman,  E.,  Anomalies  of  Visual  Acuity  in  Relation  to  Stimulus-distance,  /.  Optical  Soc.  Am.,  Vol. 

22,  285  (1932). 

Hecht,  S.,  Quantum  Relations  of  Vision,  /.  Optical  Soc.  Am.,  Vol.  32,  42  (January  1942). 
Hecht,  S.,  Relation  between  Visual  Acuity  and  Illumination,  J.  Gen.  PhysioL,  Vol.  36,  335  (1915). 
Hecht,  S.,  The  Visual  Discrimination  of  Intensity  and  the  Weber-Fechner  Law,  J.  Gen.  PhysioL,  Vol. 

7,  235-267  (1924). 
Hecht,  S.,  The  Nature  of  the  Photoreceptor  Process,  Chapter  14,  Murchinson's  Handbook  of  General 

Experimental  Psychology.     Clark  Univ.  (1934). 

Hecht,  S.,  and  E.  U.  Mintz,  Visibility  of  Single  Lines,  etc.,  J.  Gen.  PhysioL,  Vol.  22,  593  (1939). 
Hecht,  S.r  and  R.  E.  Williams,  The  Visibility  of  Monochromatic  Radiation  and  the  Absorption  Spec- 
trum of  the  Visual  Purple,  J.  Gen.  PhysioL,  VoL  5,  1-33  (1922). 
Jones,  L.  A.,  The  Fundamental  Scale  for  Pure  Hue  and  Retinal  Sensibility  to  Hue  Differences,  J. 

Optical  Soc.  Am.,  Vol.  1,  63-77  (1917). 

Helley,  K.  L.,  Color  Designations  for  Lights,  /.  Optical  Soc.  Am.,  Vol.  33,  627  (November  1943). 
Klein,  G.  S.,  Relation  between  Motion  and  Form  Acuity  in  Parafoveal  and  Peripheral  Vision  and 

Related  Phenomena,  Arch.  PsychoL,  Vol.  39,  1-69  (October  1942). 


ELECTRON   OPTICS 


14-49 


LB(ial7)!8t  N"  M"  S"  Ezperiments  on  Binocular  Vision,  Tran*  Optical  Sac.  Lwds>n,  VoL  28,  4S-I03 
1  SensitK'ities  to  C°l°r  Differences  in  Daylight,  J.  Optical  to.  4«.,  VoL  32, 

>c.  ,1m,,  VoL  34, 


M605\0ctober?9^) 


?ro?1?' 


Data  Applied  to 

-     MuMdl  CoiOT  Co.,  Baltimore  (1929). 

esolution  as  a  Function  of  Intensity  and  Exposure  Time  of  the 
Vo1'  34'  73S  December  19445. 

-  of  y-io-  *««*«  °f  Color  Vision, 


r 


^TT  nsities  of  Ugfat,  J.  Optical  Soc.  4m.,  Vol.  4,  35  . 

+L'n"  S?*11'  The  Relative  Merits  oM&d  and  Whdt«  Ugfat  of  Low  Intimity  for 
'  Efekn^ss'  ^.  O^icol  5oc.  Am.,  Vol.  34,  601  (OctoberT944). 

*»••  VoL  ^  313-322  (1938);  also  The  Shape  of 


wHr  ?•"  ^"^an  Y151^11  and  $e  Spectrum,  Sc&netf,  Vol.  101,  653  (June  29,  1945). 
Walls,  O.,  factors  m  Human  Visual  Resolution,  J.  Optical  Soc.  Am.,  V<^  33,  489  (September  19431. 
OsST'          Indlrect  Vjsusl  Acuity,  Zeitechrift  /fir  Ps^,  u^f  PA^u^.  ^.  ^innwswg.,  Vc^.  7,  172 


N  L 


ELECTRON  OPTICS 

By  D.  W.  Epstein 

GENERAL  CONCEPTS.  Electron  optics  has  become  a  branch  of  applied  physics. 
The  term  "electron  optics"  was  chosen  because  of  the  similarity  between  the  path  of  an 
electron,  or  any  charged  particle,  moving  through  electrostatic  and  rnagnetostafcie  fields 
and  that  of  a  ray  of  light  passing  through  refracting  media.  Because  of  this  similarity, 
such  concepts  of  optics  as  lens  and  focal  length  may  be  transferred  to  electrostatic  and 
magnetostatic  fields. 

The  subject  of  optics  is  generally  divided  into:  (1)  geometrical  optics,  which  treats  only 
of  the  geometrical  relations  of  the  propagation  of  light;  and  {2}  physical  optics,  which, 
utilizing  the  wave  theory  of  light,  is  capable  of  dealing  with  any  problem  in  light.  A  simi- 
lar division  is  made  in  electron  optics.  This  chapter  will  concern  itself  exclusively  with 
non-relativistic  geometrical  electron  optics  of  static  fields;  that  is,  relativity,  wave  mechan- 
ics, and  electron  optics  of  high- 
frequency  phenomena  will  be 
excluded. 

The  analogy  between  electron 
optics  and  light  is  indicated  in 
Figs.  1  and  2.  Figure  1  shows 
the  trajectory  of  a  ray  of  light 
refracted  and  reflected  at  a 
spherical  interface  separating 
two  regions  of  different  indices 
of  refraction.  The  analogous 
electron  optical  case  is  shown  in 
Fig.  2.  Assume  that  by  some 
means  (such  as  two  closely 
spaced  meshes  at  different  elec- 
trostatic potentials)  a  space  in 
vacuum  is  divided  into  two 

regions  by  a  spherieocylmdrical  .  fr.^^f     ±  *      J-D^^J* 

<mrffloP  as  shown  in  Fit?  *>  the  FIG.  1.  Trajectory  of  a  Ray  of  Light  Refracted  and  Beftecied  at 
surface  as  shown  m  Jfig.  -,  tne  &  Splierical  interface  Separating  Two  Regions  of  Different 
region  to  the  left  of  the  spherical  Indices  of  Refraction 

surface  being  at  the  electrostatic 

potential  Vi  and  the  region  to  the  right  of  the  spherical  surface  being  at  the  electrostatic 
potential  Vz-  An  electron  emitted  with  zero  initial  velocity  by  a  cathode  (located  to  the 
left  of  the  Vi  region)  at  zero  potential  will  move  in  the  Vi  region  at  a  constant  velocity  f  i 
given  by 


v^'1  meters  P61  sec 


kg  is 


where  e  =  1.59  X  10~19  coulomb  is  the  charge  of  the  electron  and  m  =  9.04  X 
its  mass.  As  long  as  the  electron  stays  in  the  region  of  constant  potential  Vi  there  is  no 
electrostatic  force  acting  on  it  and  it  will  move  in  a  straight  line,  say  OP  in  Fig.  2;  this 
corresponds  to  the  law  of  rectilinear  propagation  in  light  optics. 


14-50 


OPTICS 


When  the  electron  crosses  the  spherical  surface  at  P  its  velocity  is  changed  to  02  cor- 
responding to  F2,  i.e.,  to  z>2  =  5.95  X  105  Vvl  meters  per  second.  The  force  being  normal 
to  the  surface,  only  the  component  of  velocity  VR  normal  to  the  surface  will  change;  the 
tangential  component  of  velocity  VT  will  be  the  same  on  both  sides  of  the  surface.  It  thus 
follows  (see  Pig.  2)  that 

VT  =  0i  sin  i  —  #2  sin  r  \«) 


where  i  and  r  are  the  angles  of  incidence  and  refraction,  respectively, 
also  be  written 

Vvl  sin  i  —  Vr^i  sin  r 
or 


Equation  (2)  may 


(3) 


so  if  NI  and  #2  are  identified  as  the  indices  of  refraction  of  the  left  region  and  right  region 
respectively  and  N  as  the  relative  index  of  refraction  then  eq.  (3)  becomes  the  well-known 

law  of  refraction.  The  focusing 
properties  of  a  spherical  refract- 
ing surface  then  follow  as  indi- 
cated in  Fig.  2. 

If  F2  <  Vi,  then  e(V2  -  Fi) 
is  negative,  and  if  in  absolute 
magnitude  it  is  greater  than 
!/2  TO  (t»i  cos  i)2 — the  part  of 
kinetic  energy  of  the  electron 
corresponding  to  the  normal 
component  of  its  velocity — then 
the  electron  will  be  shot  back 
from  the  surface  with  its  nor- 
mal velocity  component  re- 
versed. The  path  of  the  re- 
flected electron  will  make  with 
the  normal  to  the  surface  the 
same  angle  i  which  its  path 
made  on  incidence  (see  Fig.  2). 
Thus  the  law  of  reflection  also 
holds  in  electron  optics. 

The  different  rays  of  a  beam 
of  light  do  not  affect  one  an- 
other. The  various  electrons 
in  an  electron  beam  repel  one 
another;  this,  of  course,  is  the 
well-known  effect  of  space 
charge.  However,  for  low  elec- 


I  VP-VI  1  >  f- OT  tei cos 

FIG.  2.     Electron  Optical  Analogy  of  Pig.  1 


tron  beam  intensities  the  effect  of  space  charge  is  negligible,  and  it  will  be  so  assumed  in 
what  follows. 

It  is  generally  customary  to  deduce  the  laws  of  geometrical  optics  especially  for  non- 
homogeneous  media  from  Fermat's  principle  of  shortest  optical  path. 

This  principle  states  that  the  path  of  a  ray  of  light  from  a  point  A  to  a  point  B  is  always 
such  as  to  make  the  integral  an  extremum  (usually  a  minimum)  with  respect  to  all  neigh- 
boring paths  for  rays  of  the  same  frequency.  The  principle  is  usually  stated  as 


N  ds  =  0 


(4) 


where  N,  the  index  of  refraction,  may  be  a  function  of  direction  as  well  as  position  and  ds 
is  an  element  of  path  length. 

In  particle  dynamics  there  is  the  similar  principle  of  least  action  stating  that 


r 

d  f 

J  A 


pds  =  0 


(5) 


where  p  is  the  momentum  of  the  particle.  A  comparison  of  eqs.  (4)  and  (5)  shows  that 
an  electron  in  an  electrostatic  and  magnetostatic  field  will  follow  the  same  trajectory  as 
light  would  if  the  index  of  refraction  at  every  point  were  made  proportional  to  the  momen- 


ELECTROSTATIC  LENSES  14-51 

turn  of  the  electron  at  the  point.  The  momentum  of  an  electron  moving  in  an  electrostatic 
and  magnetostatic  field  is 

p  =  me  —  e  A  cos  6 

where  0  is  the  angle  between  the  direction  of  motion  of  the  electron  and  the  direction  of 
magnetic  vector  potential  A.  The  index  of  refraction  for  an  electron  moving  in  a  com- 
bined electrostatic  and  magnetostatic  field  is 

N  =  k  \v  -  —  A  cos  0  (6) 

Equation  (6)  shows  that  owing  to  the  magnetostatic  field  A"  is  a  function  not  only  of  the 
position  of  the  electron  but  also  of  its  direction  of  motion  so  that  combined  electrostatic 
and  magnetostatic  fields  behave  like  non-homogeneous  anisotropic  media. 

It  was  noted  in  Fig.  2  that  the  index  of  refraction  for  an  electron  at  a  point  in  an  electro- 
static field  is  proportional  to  its  speed  at  the  point,  i.e.,  -V  —  Jb?,  but  it  must  not  be  assumed 
that  the  index  of  refraction  in  a  magnetostatic  field  is  ke/mA  cos  6  but  rather 

k     tJ0  -  —  A  cos  B\  (7) 

where  VQ  is  the  constant  speed  with  which  the  electron  moves  through  the  magnetic  field. 

14.  ELECTROSTATIC  LENSES 

The  electrostatic  lens  of  Fig.  2  is  an  example  of  a  relatively  impractical  case.  Practical 
electrostatic  lenses  are  generally  formed  by  the  application  of  electrostatic  potentials  to 
axial  symmetric  electrodes.  Figure  3  shows  a  cross-section  through  the  axis  of  such  an 
electron  focusing  system.  The  double  lines  represent  cylindrical,  hollow  conductors  at 
the  potentials  V\  and  V*\  the  single 

lines  represent  the   equipotential  g     8000080 

surfaces  in  the  space  (vacuum)  be-  £     £i  2  2  £*  ™  S  S 

tween  the  electrodes.    From  eq.  (3)  \    \   \       /  /  /  7 

it  follows  that  each  equipotential 
surface  represents  a  surface  of  con- 
stant index  of  refraction.    Here  are 
shown  only  a  few  of  the  equipoten- 
tial surfaces;  actually,  of  course, 
there  is  an  infinite  series  of  equi- 
potential surfaces  having  a  com- 
mon axis.    This  electron  focusing     FIG.  3.    Equipotential  Line  Plot  in  Charge-free  Space  Due  to 
system   may,   therefore,    be    con-          Potentials  Applie4  to  Coaxial  Cylindrical  Electrodes 
sidered  as  a  very  large  number  of 

coaxial  refracting  surfaces.  Most  optical  systems  for  light  consist  of  a  series  of  spherical 
refracting  surfaces  having  a  common  axis  of  symmetry  called  the  optic  axis.  For  light, 
the  optical  systems  are  usually  such  that  the  index  of  refraction  changes  abruptly  as  light 
passes  from  one  medium  to  the  other.  In  electron  optics,  the  index  of  refraction  is  a  con- 
tinuous function  of  position.  Optically  speaking,  this  means  that  an  electrostatic  field 
constitutes  an  isotropic  non-homogeneous  medium  for  electrons,  corresponding  to  a  me- 
dium of  continuously  variable  density  for  light  rays. 

The  potential  distribution  V(r,  z) — the  potential  at  a  radial  distance  r  from  the  axis 
and  an  axial  distance  z  from  the  origin — in  charge-free  space  due  to  potentials  applied  to 
axial  symmetric  electrodes  is  given  in  cylindrical  coordinates  by  the  Laplace  equation 


<fr»  r       dr  dz* 

subject  to  the  boundary  conditions  that  V(r,  z)  at  the  electrodes  assumes  the  values  of 
the  potentials  applied  to  the  electrodes.  Except  for  some  special  cases  it  has  not  been 
possible  to  obtain  mathematical  solutions  subject  to  the  actually  existing  boundary  condi- 
tions. The  solutions  are  generally  obtained  experimentally  by  means  of  an  electrolytic 
tank.  The  potential  distribution  of  Fig.  3  was  thus  determined. 

The  equations  of  motion  of  an  electron  moving  in  a  meridian  plane,  i.e.,  a  plane  contain- 
ing the  axis,  are 


(r,  a) 

dr 


14-52  OPTICS 

The  trajectory  of  a  meridional  electron  traversing  an  axially  symmetric  electrostatic  field 
V(r,  z)  and  with  a  velocity  v  —  \/2(e/m)V(r,  z),  namely,  an  electron  emitted  with  zero 
velocity  from  a  cathode  at  zero  potential,  is  given  by  the  differential  equation 


dV(r,  z)  dr 


d2  2V(r,  z)  62      d2  27(r,  2)  dr 

The  distribution  of  potential  in  space  is  uniquely  determined  if  the  distribution  along 
the  axis  together  with  its  even  derivatives  are  known.    Thus 


V(r,  2)  =  7(0,  2)  -       7"(0,  z)  +          V^CO,  *)  +  •••  (10) 

where  V(Q,  z)  is  the  distribution  of  potential  along  the  axis,  "F"(0,  z)  is  the  second  deriva- 
tive of  axial  potential  with  respect  to  2,  YIV(0,  2)  is  the  fourth  derivative,  etc. 

It  may  be  shown  by  using  eq.  (10)  that  the  equipotential  surfaces  in  the  neighborhood 
of  the  axis  are  hyperboloids  with  a  radius  of  curvature  of 


This  shows  that,  in  electron  optics,  index  of  refraction  and  curvature  cannot  be  varied 
independently  as  they  can  in  light  optics.  Consequently,  the  correction  of  some  lens 
aberrations  is  extremely  difficult  if  not  impossible  in  electron  optics. 

An  optical  system  is  usually  described  in  terms  of  paraxial  or  first-order  imagery.  Actual 
imagery  departs  from  paraxial  imagery.  Such  departures  are  described  as  aberrations. 

The  focusing  action  of  an  electrostatic  field  is  similarly  described  to  a  first  approxima- 
tion by  considering  only  paraxial  electrons.  Paraxial  electrons  are  characterized  by  the 
fact  that  in  calculating  their  paths  it  is  assumed  that  their  distances  from  the  axis,  r,  and 
their  inclination  toward  the  axis,  dr/dz,  are  so  small  that  the  second  and  higher  powers  of 
r  and  dr/dz  are  negligible. 

The  differential  equation  for  the  trajectory  traversed  by  a  paraxial  electron  becomes 
from  eq.  (9) 

+jr" 

iv»r-o  (13) 

4 

where  Vt  V,  and  V  are  the  axial  distribution  of  potential  V(Q,  2)  and  its  first  two  deriva- 
tives with  respect  to  z  respectively.  Equation  (12)  or  its  equivalent  eq.  (13)  may  be  taken 
as  the  fundamental  equation  of  paraxial  electron  optics  of  axially  symmetric  electrostatic 
fields. 

The  general  solution  of  the  second-order  differential  equation  (12)  or  (13)  may  be  written 


r(2)  -  cin(2)  +  C2r2(2)  (14) 

where  r\(z)  and  r2(2)  are  any  two  linearly  independent  solutions  and  Ci  and  c2  are  arbitrary 
constants.  Equation  (14)  states  that  the  trajectory  of  any  paraxial  electron  is  simply  the 
linear  combination  of  two  independent  trajectories.  Hence  the  complete  paraxial  focus- 
ing action  of  an  axially  symmetric  electrostatic  field  is  determined  by  calculating  the 
trajectories  of  only  two  electrons.  The  trajectories  of  two  electrons  entering  the  lens 
parallel  to  the  axis  from  the  object  and  image  sides  are  chosen  as  the  two  solutions  7-1(2) 
and  ri(z)  and  are  called  the  two  fundamental  trajectories.  These  trajectories  determine  the 
location  of  the  cardinal  points  of  the  focusing  system,  i.e.,  the  location  of  the  focal  and 
principal  points,  and  thus  the  focusing  action  of  the  lens. 

Referring  to  Fig.  4  let  Si  and  8%  be  two  equipotential  surfaces  such  that  the  space  to 
the  left  of  Si  is  equipotential  and  is  at  potential  Vi  and  the  space  to  the  right  of  S%  is  equi- 
potential and  at  the  potential  V^  The  potential  in  the  region  between  Si  and  £3  varies 
continuously,  as  shown  in  Fig.  3.  Then  the  paraxial  electron  moving  parallel  to  the  axis  in 
the  equipotential  space  to  the  left  of  Si  will  follow  the  trajectory  7-1(2)  and  after  passing 
through  the  focusing  system  will  move  in  a  direction  inclined  at  an  angle  to  the  axis  and 
will  pass  through  the  axial  point  -FV  All  paraxial  electrons  moving  parallel  to  the  axis  in 
the  Vi  or  object  space  will  pass  through  F%.  The  point  ^2  is  the  second  focal  point.  The 
plane  passing  through  the  second  focal  point  and  perpendicular  to  the  axis  of  symmetry  is 
the  second  focal  plane. 


ELECTEOSTATIC  LENSES 


14-53 


The  plane  perpendicular  to  the  axis  and  passing  through  the  point  of  intersection  of  the 
original  and  final  directions  of  motion  of  the  electron  is  the  second  principal  plane.  The 
point  of  intersection  H 2  between  the  second  principal  plane  and  the  axis  is  the  second  prin- 
cipal point.  The  distance  H*FZ  denoted  by  /*  is  the  second  focal  length. 

Similarly,  a  paraxial  electron  moving  parallel  to  the  axis  in  the  T%  or  image  space  will 
after  passing  through  the  focusing  system  move  in  a  direction  inclined  10  the  axis  and  will 


:ond  Focal 


.Second  Focal  Point 


FIG.  4.     Fundamental  Trajectories  and  Cardinal  Points  of  an  Electrostatic  Lens 


pass  through  the  point  FI  in  the  object  space.  FI  is  the  first  focal  point.  Fi  may  also  be 
considered  that  axial  point  in  the  object  space  from  which  all  electrons,  after  passing 
through  the  focusing  system,  are  parallel  to  the  axis  in  the  image  space. 

The  plane  perpendicular  to  the  axis  of  symmetry  and  passing  through  the  first  focal 
point  is  the  first  focal  plane.  The  plane  perpendicular  to  the  axis  and  passing  through  the 
point  of  intersection  of  the  original  and  final  directions  of  motion  of  the  electron  is  the 
first  principal  plane.  The  point  of  intersection,  J5Fi,  of  the  first  principal  plane  and  the 
axis  is  the  first  principal  point. 

It  is  to  be  noted  that  in  Fig.  4  the  principal  planes  are  crossed.  This  is  a  characteristic 
of  lenses  having  indices  of  refraction  different  on  the  two  sides. 

In  Fig.  5  let  AiBi  be  an  object  (say,  an  aperture  through  which  electrons  are  passing) 
located  in  the  equipotential  region  Vi.  Then  a  paraxial  electron  coming  from  AI  and 
moving  parallel  to  the  optic  axis  will  after  passing  through  the  lens  go  in  the  direction 
F*A*.  A  paraxial  electron  issuing  from  A\  in  the  direction  AiFi  will  after  passing  through 
the  lens  move  parallel  to  the  optic  axis  and  will  intersect  the  trajectory  of  the  other  electron 
at  Az-  Similarly  the  trajectories  of  all  paraxial  electrons  coming  from  A\  will  intersect  at 
A  2,  the  image  of  A\.  The  same  is  true  of  every  point  on  AiBi,  and  so  the  inverted  image 


FIG.  5.     Image  Formation  in  a  Direct  Bipotential  Electrostatic  Lens 

j,  is  obtained.    The  ratio  AzB»/AiBi  or  h*/hi  gives  the  magnification.    The  electron 

image  of  AiBi  becomes  visible  if  a  fluorescent  screen  is  placed  in  the  plane  of  A^BZ. 
From  eq.  (12)  or  (13)  and  Fig.  5  it  may  be  shown  that  the  following  relations  hold: 

&  -      \m  (15) 

(16) 


/i  + 


•+; 


/* 


+  </i  - 


tfc  -  . 


(17) 


14-54  OPTICS 


... 

7F"  LP  +  C/i  —  - 

gi 


(20) 


2  „  (F2  -  /a)  -  fz(m  -  1)  (21) 

The  complete  paraxial  focusing  action  of  an  electrostatic  lens  is  determined  by  means  of 
the  above  relations  when  the  positions  of  the  cardinal  points  or  /i,  /2,  -Fit  and  F%  are  known. 

Electrostatic  lenses  are  classified  according  to  (a)  electrode  symmetry,  (6)  thickness, 
and  (c)  the  potentials  on  the  sides  of  the  lens. 

Spherical  lenses  are  formed  by  applying  different  voltages  to  two  or  more  electrodes 
having  axial  symmetry  such  as  apertures  and  cylindrical  and  conical  tubes.  Spherical 
lenses  are  used  in  cathode-ray  tubes,  television  pick-up  tubes,  and  electron  microscopes. 

Cylindrical  lenses  are  formed  by  applying  different  voltages  to  two  or  more  pairs  of 
electrodes  having  a  plane  of  symmetry  such  as  pairs  of  wires  or  pairs  of  strips.  Cylin- 
drical lenses  are  used  in  some  receiving  and  transmitting  tubes. 

Thick  lenses  are  characterized  by  the  fact  that  the  axial  extension  of  the  electrostatic 
or  refracting  field  is  of  the  same  order  of  magnitude  as  or  even  larger  than  the  focal  length. 
It  is  necessary  to  know  the  positions  of  all  four  cardinal  points  in  order  to  determine  the 
focusing  action  of  a  "thick  lens.'* 

Thin  lenses  are  characterized  by  the  fact  that  the  axial  extension  of  the  refracting 
field  is  negligible  compared  with  the  focal  length.  In  a  thin  lens  the  principal  planes 
coincide,  and  the  focusing  action  of  the  lens  is  determined  by  its  location  and  focal  lengths. 
For  estimating  purposes  it  is  sufficiently  accurate  to  consider  most  lenses  "thin." 

The  focal  lengths  of  a  "thin"  spherical  lens  may  be  calculated  from  the  relations 


(22) 


where  V\  and  Vz  are  the  potentials  of  the  equipotential  regions  on  the  two  sides  of  the 
lens,  y  and  V  the  axial  distribution  of  potential  and  its  first  derivative  with  respect  to  z, 
and  the  integral  is  taken  over  the  axial  extension  of  the  field. 

Equations  (22)  also  apply  to  "thin"  cylindrical  lenses,  if  the  numerical  factor  3/16  is 
replaced  by  1/2- 

Unipotential  lenses  are  characterized  by  having  identical  equipotential  regions  on  the 
object  and  image  sides,  Vi  —  Vz,  and  hence  /i  =  —ft.  Figure  6  shows  a  few  electrode 
arrangements  and  axial  distributions  of  potential  of  unipotential  lenses.  Figures  7  and  8  * 
give  the  focal  length  of  several  thin  unipotential  lenses  as  a  function  of  Ve/Vo.  The  dotted 
curves  of  Fig.  8  give  the  focal  length  of  a  unipotential  lens  where  the  central  aperture  has 
been  replaced  by  a  fine  metal  screen  or  an  electron-permeable  conducting  membrane.  It 
should  be  noted  that  this  lens  is  divergent,  i.e.,  fz  is  negative,  when  Ve/Vv  is  less  than  unity. 
In  the  case  of  thin  unipotential  lenses  eqs.  (15)  to  (21)  simplify  to 

A  -  -h  (150 

XiXz  -  -/a2  (160 

i-i  =  ^  (180 

Q.       P      h 

»  =  r  =  !-=-^  =  !  =  ii  =  -  (wo 


1        3  fVi\X  r*  (V'\*  . 

7*~iS(v)  L  (v)dz 

1  3  fV*\X  /•&  (V'\  , 

*--ieU;  L  (v)  dz 


p  =  -f*(r^r)  (20° 

5  =  -Mm  -  1)  (210 

*  Figures  7  and  8  ware  plotted  from  data  calculated  by  Dr.  E.  G.  RambergW  RCA  Laboratories. 


ELECTROSTATIC  LENSES 


14-55 


Vi      V? 


Vl 


FIG.  6.    Electrode  Arrangements  and  Axial  Distribution  of  Potential  of  Some  Unipotential  Lanses 


14 
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Vg  _  Voltage  Applied  to  Central  Electrode 
Vfl       Voltage  Applied  to  Outer  Elictrodes 

FIG.  7.    Variation  of  the  Focal  Length  (Measured  as  the  Number  of  Diameters  of  the  Central  Aperture) 
with  the  Ratio  of  the  Voltages  Applied  to  the  Central  and  Outer  Electrodes 


O.S   1 


1,4       1.8       2.2      2.6       3,0      3.4 
,  Voltage  Applied  to  Central  Electrode 
=  Voltage  Applied  to  Outer  Etecirodes 


FIG.  8.    Variation  of  the  Focal  Length  (Measured  as  the  Number  of  Diameters  of  the  Outer  Electrodes) 
with  the  Ratio  of  the  Voltages  Applied  to  the  Central  and  Outer  Electrodes.    The  solid  curves  apply  to 
system  (a)  with  a  small  central  aperture.     The  dotted  curves  apply  to  system  {&)  with  a  central  elec- 
trode consisting  of  a  fine  metal  mesh  or  an  electron-penaeable  conducting  membrane. 


14-56 


OPTICS 


Bipotential  lenses  are  characterized  by  having  different  equipotential  regions  on  the 
object  and  image  sides,  or  Vi  7^  V*.  In  the  direct  bipotential  lens  the  potential  of  the  image 
space  is  greater  than  that  of  the  object  space  (Vz  >  7i).  In  the  inverted  bipotential  lens 
V-2  <  V\.  Figure  9  shows  a  few  electrode  arrangements  and  axial  distributions  of  poten- 
tial of  some  bipotential  lenses.  Figures  10a  to  lOe  give  the  focal  lengths  and  positions  of 


Vi 


FIG.  9.    Electrode  Arrangements  and  Axial  Distribution  of  Potential  of  Some  Bipotential  Lenses 

the  focal  points  (thus  giving  the  positions  of  the  four  cardinal  points,  see  Fig.  5)  as  a  func- 
tion of  Vz/Vi  for  5  different  diameter  ratios  d%/d\.  Figure  11  shows  /i,  /2,  FI,  and  F%  as  a 
function  of  dz/di  for  V%/Vi  —  5. 

The  focusing  characteristics  of  a  "thick"  bipotential  lens  are  given  by  eqs.  (15)  to  (21). 
Their  use  in  conjunction  with  Fig.  lOc  will  be  illustrated  by  the  following  simple  example 
(see  Fig.  5  for  sign  convention).  Given  the  lens  with  d^/d\  =  1.5,  V%/Vr~  5,  then,  from 
Fig.  lOc,  /i  -  -2.1dlf  CFi  -  /O  =  -l.ldi,  h  =  4.7<*i,  and  (F2  -  /a)  =  -1.8^.  It  is 
desired  to  obtain  a  real  image  of  an  object  with  a  magnification  m  =  —  5  (since  a  real 
image  of  an  object  is  inverted).  Then  by  eqs.  (20)  and  (21) 


-f  4.7di(-5  -  1) 

and  thus  the  object  is  located  at  a  distance  of  S.Gdi  to  the  left  of  the  cylinder  ends  and  the 
image  26.4^i  to  the  right  of  the  cylinder  ends. 


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G.  10.    (See  facing  page.) 


ELECTROSTATIC  LENSES 


14-67 


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FIG.  10.    Variation  of  Focal  Lengths,  /,  a&d  Distance  of  Focal  Poinfes  f rom  End 1  oj ^Oygnder/- 
as  the  Number  of  Diameters  of  the  Cylinder  on  the  Left),  with^the  Ratao  of  the  Voltages 

Eight  and  Left  Cylinders 


14-58  OPTICS 

In  the  case  of  a  "thin"  bipotential  lens  eqs.  (18)  to  (21)  simplify  to 


-f  a)       f* 


(18*) 


m-  \ 


fl  +  a 


(20") 

q  -  -a  -  Mm  -  1)  (21") 

where  a  is  the  distance  between  the  position  of  the  equivalent  thin  lens  and  cylinder  ends 
as  indicated  in  Fig.  5. 

Bipotential  lenses  are  most  generally  used  in  cathode-ray  tubes. 

Cathode   qGrfd     fAnode 


0*0.2V     +Y 


00  V 


1234. 
ds__   Diameter  of  Right  Cylinder 
di~"   Diameter  of  Left  Cylinder 

FIG.  11.  Variation  of  Focal  Lengths,  /T  and  Distances 
of  Focal  Points  from  End  of  Cylinders,  jP  (Measured  as 
the  Number  of  Diameters  of  Cylinder  at  Left),  -with  the 
Ratio  of  Diameters  of  the  Bight  and  Left  Cylinders 


0-0.2V 


FIG.  12.  Electrode  Arrangement  and 
Approximate  Distributions  of  Po- 
tential (for  Three  Values  of  Grid  Bias) 
of  One  of  the  Most  Commonly  Used 
Cathode  Lenses 


Cathode  lenses  are  characterized  by  having  a  plane  of  zero  potential  (cathode)  normal 
to  the  optic  axis.  The  cardinal  points  of  a  cathode  lens  are  of  less  significance  than  in  case 
of  the  other  lenses  because  of  the  distribution  of  initial  velocities  of  the  electrons  emitted 
by  the  cathode.  Figure  12  shows  the  electrode  arrangement  and  approximate  distribution 
of  potential  of  one  of  the  most  commonly  used  cathode  lenses.  Figure  12  also  shows  the 
approximate  trajectory  of  an  electron  emitted  normal  to  the  cathode  for  grid  voltages  of 
+0.2,  0,  and  —0.2  V.  The  grid  is  so  named  since  it  is  also  used  for  controlling  the  cur- 
rent going  to  the  anode  aperture.  The  cathode  lens  is  the  most  widely  used  since  it  exists 
in  all  types  of  electronic  devices. 

Electrostatic  lenses  suffer  from  the  defects  or  aberrations  known  as  spherical  aberration, 
coma,  astigmatism,  curvature  of  field,  distortion,  and  chromatic  aberration.  They  also 
suffer  from  defects  due  to  misalignment  and  maleonstruction  of  electrodes,  space  charge, 
relativity  effect,  etc. 

Spherical  aberration  is  in  general  the  most  troublesome,  especially  in  instruments  re- 
quiring a  fine  focused  spot  (or  line)  as  in  cathode-ray  tubes,  television  pick-up  tubes,  and 
electron  microscopes.  Figure  13  shows  the  increase  in  spot  size  caused  by  the  spherical 
aberration  of  a  bipotential  lens.  The  increase  in  spot  size  is  given  as  a  function  of  the 
beam  diameter  at  the  end  of  the  first  cylinder  measured  in  terms  of  the  first-cylinder 
diameter.  Figure  14  shows  the  decrease  in  voltage  ratio  required  to  focus  a  beam  of 


MAGNETOSTATIC  LENSES 


14^59 


tK^nf ff v"    +     ?-Ti       h€  spherical  aberration  of  the  unipotential  lens  is  greater  than 
!?*„  fi      blfotentmi  k?8,  ?"*  the  equldiameter  bipoiential  lens  has  Jess  spherical  aber- 
ration than  a  lens  with  (d.Jd,}  >  I  or  <*/*)  <  1.    In  general  the  aberrations  in  electron 

lenses  are  much  more  severe  than  they  are  in 
light  lenses  (this  also  applies  to  magnetos* atie 
lenses).  This  is  primarily  due  to  the  connec- 
tion between  radius  of  curvature  and  index  of 
refraction — see  eq.  (11) — which  exists  in  elec- 
tron optics  and  not  in  light  optics. 


1500 

I 

1400 

V? 

V- 

J 

S1300 



f1 

—  =: 

—i    — 

—  —  -. 

3 

c  

-J 

!) 

4. 

3 

•51200 

<Zf> 

_i 

rr 

wnon 

di 

=  1 

b 

rt 

Si  ooo 

^  900 

) 

/ 

£  800 

/ 

g  700 

/ 

£ 

«=  600 

^ 

0> 

£  50° 

/ 

I  400 

/ 

/ 

w 
300 

/ 

200 

^ 

/ 

100 

—-  — 

^ 

.0    0.1  0.2  O.3  0.4  0.50.6  0.7  0.8  0.9  1.0 

d  _  Beam  Diameter  at  End  of  First  Cylinder 
d-L  Diameter  of  First  Cylinder 

FIG.  13.    Increase  in  Spot  Size  Caused  by  t&e 

Spherical  Aberration  of  a  Bipotential  Lens  as 

a  Function  o!  the  Beam  Diameter  at  the  End 

of  the  First  Cylinder 


0.90 

S  0.80 
a 
0.70 

S0.6O 

<3 

^0,50 
o  0.40 

i^~^4^i     i          ! 

! 

i  ^>-<^  ! 

^^s^ 

i 

*  0.30 
20.20 
>0.10 

0,00 
0. 

i 

j       :       i              l 

j 

i 

1 

! 

i    ! 

!           !    1 

0  ai  0.2  03  0,4  0.5  0.6  0.7  Q.8  0.9  1. 

FIG.  14.    Decrease  in  Focusing  Voltage  Ratio 
Caused  by  the  Spherical  Aberration  of  a  Bipo- 
tential Lens  as  a  Function  of  she  Beam  Diam- 
eter at  the  End  of  the  First  Cylinder 


15.  MAGNETOSTATIC  LENSES 

Any  axially  symmetric  magnetostatic  field  acts  as  a  "lens"  for  electrons.  Such  fields 
are  generally  created  by  passing  direct  current  through  coils  with  axial  symmetry. 

As  already  noted,  a  magnetostatic  field  is  an  electron-refracting  medium  whose  index 
of  refraction  is  a  function  not  only  of  position  but  also  of  the  direction  of  travel  of  the 
electron.  Consequently  magnetostatic  lenses  differ  in  then-  behavior  from  electrostatic 
and  ordinary  light  lenses.  In  general  the  differences  are  made  evident  by  a  rotation  of 
the  image  and  the  irreversibility  of  object  and  image.  ThusT  a  real  image,  produced  by  a 
magnetostatic  lens,  is  not  inverted  (as  it  is  in  an  electrostatic  lens),  but  is  rotated  through 
an  angle  6  relative  to  the  inverted,  image.  Similarly,  if  the  image  is  made  the  object,  then 
its  image  will  not  coincide  with  the  original  object  (as  it  would  in  an  electrostatic  lens) 
but  will  be  rotated  through  an  angle  28  relative  to  the  original  object.  Magnetostatic  focus- 
ing is  generally  accomplished  by  long  or  short  lenses.  The  image  rotation  of  a  magnetosattic 
lens  is,  for  a  given  focal  length,  reduced  as  the  extent  of  the  field  along  the  axis  is  reduced. 
Thus  the  image  rotation  is  practically  zero  for  aa  extremely  short  lens  and  is  180°  (image 
erect)  for  a  very  long  lens. 

The  long  lens  generally  extends  over  the  entire  length,  of  the  electron  beam  and  is  formed 
by  the  uniform  field  of  a  long  solenoid.  It  is  used  in  such  television  pick-up  tubes  as  the 
image  dissector,  orthicon,  and  image  orthicon.  The  uniform  field  of  the  long  lens  produces 
a  sequence  of  uniformly  spaced,  real,  erect  (rotation  180°)  images  with  unity  magnifica- 
tion of  an  object  which  is  placed  normal  to  the  field  and  which  emits  electrons  of  uniform 
speeds.  This  follows  from  the  fact  that  an  electron  injected  into  a  uniform  magnetostatic 
field,  B,  with  a  speed  v,  and  a  corresponding  voltage  V  and  at  an  angle  a  with  the  field, 
describes  a  helix,  the  radius  of  the  helix  being 


flsina       3.38  X  10~s 


(fl/m)B 


-  sin  at  meters 


(23) 


H 


-meters 


14-60 


OPTICS 


The  time  taken  by  the  electron  in  describing  1  revolution  is  * 

_  2x  3.56  X  1CT11       2.84  X  10~s 


B 


H 


(24) 


(e/m)B 

which  depends  exclusively  on  B  or  H  and  is  independent  of  V  or  ex.     The  pitch  of  the 
helix  or  the  distance  that  the  electron  has  traveled  in  time  T  is 

21.1  X  IQ~*\/V  cos  a  , 

: meters   \ 


27rtj  cos  a. 
(e/m}B 


-  meters 


(25) 


If  the  angles  a.  at  which  electrons  from  a  point  source  are  injected  into  the  field  are  small, 
then,  their  speeds  along  the  lines  of  force,  »  cos  a,  are  essentially  constant  (cos  a.  —  1) 
and  all  electrons  will  reunite  at  the  following  distances  from  the  source 


21.1  X  10"6nV7 
B 


H 


meters 


(26) 


where  n  is  an  integer.  Since  any  line  of  force  is  an  axis  of  symmetry  in  a  uniform  field, 
an  extended  electron-emitting  (or  "illuminated")  source  placed  perpendicular  to  the  field 
will  be  imaged  at  the  distances  sn  of  eq.  (26)  (see  Fig.  15) . 


Object 


[ O  O  O  Q  O  O  O  Q  O  O  O  Q  Q  O  Q  O  O  O  OOOOOOOOOOO  OOP  O  CM 

-H  HH 


FIG.  15.     Image  Formation  by  a  Uniform  Magnetostatic  Field 

The  short  magnetostatic  lens  extends  over  a  limited  region  of  the  beam  and  is  generally 
formed  by  the  non-uniform  field  of  an  iron-encased  coil  with  a  narrow  slit  in  the  iron  shell 
or  pole  pieces  (Fig.  16).  Short  lenses  are  used  in  electron  microscopes  and  television 


FIG.  16.    Short  Iron-dad  Magnetostatic  Lens  and  Approximate  Distribution  of  Axial  Flux  Density 


cathode-ray  tubes.    Except  for  some  image  rotation  the  short  lens  behaves  like  a  thin 
uni|M>tential  electrostatic  lens  so  that  its  focusing  characteristics  are  given  by  eqs.  (150 


MAGNETOSTATIC  LENSES 


14-61 


to  (21').    The  focal  length  of  a  short  lens  is  given  by 

2.2  X  1010  f 
V        Ja 
3.5  X  IO-2  t 


meters" 


meters 


(27) 


where  B  is  the  axial  component  of  the  flux  density  along  the  axis  of  the  lens,  and  V  is  the 
potential  of  the  equipotential  region  in  which  the  lens  is  located. 
The  image  rotation  is  given  by 


8  m 


radian* 


=  0.19  r*> 

~   */vJa  ' 


H  dz     radians 


(28) 


Integrating  eqs.  (27)  and  (28)  for  a  wire  loop  of  diameter  d  and  carrying  a  current  ATI 
results  in 


/a  8  X  W^V  V 

d       37r3(e/m)(ArJ)2  .V2!2 


10~ 


NI 

( 

'V? 


(29) 
(30) 


To  a  fair  approximation,  eq.  (29)  also  applies  to  a  short  coil  of  mean  diameter  d,  of  N  turns, 
and  carrying  current  /.    Figure  17  is  a  plot  of  eq.  (29)  giving,  graphically,  the  focal  length 


4   6  S  1O    20    4O  60  80 1OO   2OO    400  600  1000 


FIG.  17.     The  Focal  Length  of  a  Short  Coil  as  a  Function  of  Ampere-turns  and  Beam  Voltage 

as  a  function  of  voltage  and  ampere-turns.    Figure  17  may  be  used,  for  estimating  pur- 
poses,  in  the  case  of  an  iron-encased  coil  if  the  ampere-turns  are  small  and  if  d  is  taken  as 


14-62 


OPTICS 


the  clear  diameter  of  the  pole  pieces.    Since  the  iron-clad  coil  concentrates  the  field  more 
than  the  air  coil,  the  iron  coil  produces  the  smaller  image  rotation. 

Besides  the  usual  5  third-order  aberrations  of  spherical  aberration,  coma,  astigmatism, 
distortion,  and  curvature  of  field,  the  images  produced  by  magnetostatic  lenses  usually 
suffer  from  three  other  aberrations.  These  are  generally  known  as  anisotropic  distortion, 
anisotropic  curvature  of  field,  and  anisotropic  coma.  Anisotropic  distortion  is  often 
called  the  "S  effect/'  since  this  aberration  distorts  a  straight  radial  line  on  the  object  into 
an  elongated  letter  S  on  the  image. 


16.  ELECTRON  PRISMS 

A  uniform  electrostatic  field  and  a  uniform  magnetostatic  field  constitute  the  two  basic 
types  of  electron  prisms.  Electron  prisms  are  used  for  deviating  (deflecting)  beams  of 
electrons  of  uniform  speed  as  in  cathode-ray  tubes  and  pick-up  tubes,  and  for  dispersing  a 
beam  of  charged  particles  so  as  to  separate  particles  of  differing  mass  or  speed  as  in  the 
ion  trap  of  a  cathode-ray  tube  and  mass  spectrograph. 

Electrostatic  prisms  are  generally  formed  by  the  approximately  uniform  field  between 
the  charged  plates  of  a  condenser  (see  Fig.  18) . 


FIG.  18.    Deviation  (Deflection)  of  an  Electron  Beam  by  an  Electrostatic  Prism  (Deflecting  Plates) 

A  charged  particle  moving  initially  perpendicularly  to  a  uniform  electrostatic  field,  13 , 
with  a  velocity  v  (corresponding  to  a  potential  drop  V},  follows  a  parabolic  trajectory  while 
in  the  field.  On  leaving  the  field  the  charged  particle  is  deviated  into  the  direction  of  the 
field  (see  Fig.  18)  through  the  angle  a  given  by 


tan  ce.  • 


__ 
2V  '' 


IVd 
''  2dV 


(31) 


where  I  is  the  extent  of  the  field  or  approximately  the  length  of  the  plates,  d  the  distance 
between  the  plates,  and  Vd  is  the  difference  in  potential  between  the  plates.  The  apparent 
center  of  deflection  is  located  approximately  at  1/2  from  the  ends  of  the  plates,  and  the 
amount  of  deflection  at  the  distance  L  from  the  center  of  the  plates  is 


ILVd 


(32) 


Magnetostatic  prisms  are  generally  formed  by  the  approximately  uniform  field  between 
two  current-carrying  coils  or  pole  pieces  of  a  magnet. 


Ifca,  1$.    DeTiati©D.  (BefleotioiL)  of  an  Electron  Beam  by  a  Magnetostatic  Prism  (Deflecting  Coils) 


REFERENCES  14-63 

A  charged  particle  moving  initially  perpendicularly  to  a  uniform  magnetostatic  field 
with  the  velocity  t>  follows  a  circular  trajectory  in  a  plane  at  right  angles  to  the  field.  On 
leaving  the  field  the  charged  particle  is  deviated  in  a  direction  perpendicular  to  the  field 
(see  Fig.  19)  through  the  angle  a  given  by 

\/7]mlB 
sin  a.  =  -  _  (33) 

Vzv 

which  becomes  in  the  case  of  electrons 

sin  a  =  2.97  X  105  -^L  (34) 

vv 

The  apparent  center  of  deflection  is  located  approximately  at  the  center  of  the  field,  and 
the  amount  of  deflection  at  the  distance  L  from  the  center  of  the  field  is 

IT  5 
h  -  L  tan  a  S  2.97  X  105  -^  (35) 


17.  GENERAL  THEOREMS  ON  ELECTRON  OPTICAL  SYSTEMS 

Electron  optical  systems  generally  consist  of  a  cathode  lens  and  one  or  more  electrostatic 
and/or  magnetostatic  lenses  and  electron  prisms  (deflecting  plates  or  coils). 

The  following  general  theorem  applies  to  an  electron  optical  system  using  electrostatic 
and  magnetostatic  elements.  The  trajectory  of  a  charged  particle  remains  similar,  as  long 
as  the  quantity  (e/m}(BzLz/V]  is  kept  unchanged.  Thus,  if  the  voltages  on  all  the  elec- 
trodes are  increased  by  a  constant  factor  n,  it  is  necessary  to  increase  B  by  the  factor  Vn 
in  order  to  keep  the  trajectory  the  same.  If  all  the  linear  dimensions  (L)  of  the  system, 
i.e.,  of  all  the  electrodes,  coils,  object  distance,  image  distance,  etc.,  are  increased  by  the 
factor  n,  it  is  necessary  either  to  increase  all  the  voltages  by  n?  or  to  decrease  B  to  l/n 
in  order  to  keep  the  trajectory  similar. 

The  following  theorems  apply  to  any  purely  electrostatic  electron  optical  system: 

1.  The  trajectory  is  independent  of  e/m.    Hence  all  like  charged  particles  emitted  by  a 
cathode  will  be  identically  focused  and  deflected  by  any  electrostatic  optical  system. 

2.  The  trajectory  of  any  charged  particle  emitted  by  a  cathode  is  unchanged  if  the 
voltages  on  all  the  electrodes  are  increased  by  the  same  factor. 

3.  If  all  the  linear  dimensions  of  all  the  electrodes  are  increased  by  a  constant  factor  n, 
the  trajectory  remains  unchanged  (if  measured  in  units  n  times  larger). 

All  electron  lenses  which  are  free  of  space  charge  and  conductors  within  the  field  of  the 
lens  and  are  bounded  by  uniform  potential  regions  on  both  sides  are  always  convergent, 
i.e.,  will  form  a  real  image  of  an  object  located  beyond  the  focal  point  of  the  lens. 

The  maximum  current  density,  /a,  obtainable  in  an  electron  spot  or  image,  regardless 
of  the  electron  optical  system  employed,  is  given  by 


where  J"i  is  the  specific  emission  of  the  cathode,  kT  is  the  initial  kinetic  energy  of  an  elec- 
tron emitted  by  a  thermionic  cathode  at  the  absolute  temperature  T,k  =  1.38  X  10  "^  erg 
per  degree,  eVz  +  kT  is  the  final  kinetic  energy,  and  az  is  the  angle  of  maximum  converg- 
ence of  the  electron  trajectory  at  the  spot  or  image. 

Equation  (36)  may  be  deduced  from  the  fact  that,  for  a  perfect  optical  system  which 
accepts  and  focuses  all  the  electron  current  emitted  by  a  thermionic  cathode,  J*  =  Ji/m* 

(m  =  magnification),  and  m  =  —  Sm  ai  (see  eq.  [41],  p.  14-08),  where  DI  and  t?2  are  the  ini- 

$2  sin  02 

tial  and  final  velocities  of  an  electron  and  «i(=  90°)  is  the  angle  of  maximum  convergence 
at  the  cathode. 

REFERENCES 

1.  Bruche  and  Scherzer,  Geometrische  Elektronenoptik.    Julius  Springer,  Berlin  (1934). 

2.  Maloff  and  Epstein,  Electron  Optics  in  Television.    McGraw-Hill  (193S). 

3.  Meyers,  L.  M.?  Electron  Optics.    Chapman  <fc  Hall,  London  (1939). 

4.  Zworykin,  V.  K.,  et  aL,  Electron  Optics  and  the  Electron  Microscope,     John  Wiley  (1945). 


SECTION  15 
ELECTRO-OPTICAL  DEVICES 


PHOTORESPONSIVE  DEVICES 
AST.  BT  HERBERT  E.  IVES  PAGE 

1.  Radiation,    Properties,    and    Means    of 

Detection 02 

2.  Thermal  Devices 03 

3.  Photoemissive  Cells 04 

4.  Photoconductive  Cells 11 

5.  Barrier  Photocells 13 

6.  Photovoltaic  Cells 15 

7.  Choice  of  Cells  for  Various  Purposes 15 

TELEVISION  PICK-UP  TUBES 

BT   V.   K.    ZWOBTKIN  AND 

E.  G.  RAMBEBG 

8-  Requirements 19 

9.  The  Image  Dissector 19 

iQ.  The  Iconoscope . 21 

11.  The  Monoscope 25 

12.  The  Orthicon 26 

13.  The  Image  Orthicon 27 

14.  Fields  of  Application  of  Pick-up  Tubes  29 


LUMINESCENT  AND  TENEBRESCENT 
MATERIALS 


AST.  *      -  PAGE 

15.  Preparation  and  Notation  of  Phosphors  32 

16.  Mechanisms  of  Phosphors  .............  34 

17.  Mechanisms  of  Scotophors  ............  36 

IS.  Specific  Characteristics  of  Useful  Phos- 

phors and  Scotophors  ...............  37 

CATHODE-RAY  TUBES 

BT    L.    E,   SWEBLTTND 

19.  Electron  Gun  ........................  41 

20.  Bulbs  for  Cathode-ray  Tubes  ..........  43 

21.  Characteristics  of  the  Image  ...........  44 

22.  Television     Cathode-ray    Reproduction 

Tubes  ............................  46 

23.  Oscillograph-type  Cathode-ray  Tubes.  .  .  47 

24.  Cathode-ray-tube      Displays      (by     T. 

SOUJEB)  ...............  ...-  ......  49 


15-01 


ELECTRO-OPTICAL  DEVICES 
PHOTORESPONSIVE  DEVICES 

By  Herbert  E.  Ives 

1.  RADIATION,  PROPERTIES,  AND  MEANS  OF  DETECTION 

Photoelectric  tubes,  or  cells,  belong  to  the  general  class  of  instruments  responsive  to 
radiant  energy,  or  radiation.  An  intelligent  use  of  such  instruments  demands  a  knowledge 
of  the  more  important  characteristics  of  radiation,  which  may  be  summarized  as  follows: 
Radiant  energy  consists  of  electromagnetic  waves,  of  a  vast  range  of  wavelengths,  extend- 
ing from  radio  waves,  many  meters  long,  to  x-rays  and  y-rays  less  than  a  millionth  of  a 
millimeter  in  length  (see  Section  11,  article  5),  The  portion  of  this  extended  spectrum 
which  is  effective  in  photoelectric  tubes  and  similar  devices  is  confined  to  a  small  region 
centering  around  that  group  of  wavelengths  commonly  called  light,  or  the  visible  spectrum, 
which  extends  roughly  from  0.8  micron  (1  micron  =  10~4  cm)  to  0.4  micron.  (Other  units 
of  measurement  frequently  encountered  are  the  angstrom  unit,  10"8  cm,  and  the  millimi- 
cron, 10~7  cm.)  The  adjacent  region  of  longer  wavelength  is  called  the  infrared;  that  of 
shorter  wavelength,  the  ultraviolet*  The  radiations  in  the  visible  spectrum  vary  greatly  in 
their  ability  to  produce  the  sensation  of  light,  the  relationship  between  energy  value,  or 
radiant  intensity,  and  luminous  value  being  indicated  by  the  luminosity  curve  of  the  spec- 
trum which  has  a  maximum  at  approximately  0.5  micron,  dropping  to  zero  at  the  ends  of 
the  visible  spectrum.  Radiant  energy  is  measured  in  energy  units,  namely,  ergs  or  joules; 
and  radiation  or  radiant  flux  is  measured  in  ergs  per  second  or  watts:  The  irradiation  of  a 
surface,  such  as  that  of  a  phototube,  is  measured  in  ergs  per  second  incident  per  unit  area, 
or  watts  per  unit  area.  Luminous  flux  is  radiant  flux  evaluated  according  to  its  luminosity 
(the  integral  of  the  product  of  the  radiation  spectrum  and  the  luminosity  curve  of  the 
spectrum).  Its  unit  is  the  lumen.  One  lumen  is  radiated  in  unit  solid  angle  by  a  point 
source  of  luminous  intensity  of  1  candle.  The  illumination  of  a  surface  is  measured  in 
lumens  incident  per  unit  area.  The  meter-candle  or  lux  is  an  illumination  of  1  lumen  per 
square  meter. 

All  instruments  which  respond  to  radiation  do  so  because  they  absorb  some  of  the  energy 
of  the  radiation  and  transform  it  into  some  form  susceptible  of  practical  measurement  or 
utilization.  Thermal  instruments  become  heated  by  the  incident  energy,  and  indicate 
by  changes  of  dimensions,  position,  or  the  production  of  electrical  currents.  Photoelectric 
instruments  depend  primarily  upon  the  fact  that  the  incidence  of  radiation  on  matter 
causes  the  emission  or  release  of  electrons,  setting  up  electromotive  forces  or  causing 
electrical  currents  to  flow. 

CLASSIFICATION"  OF  PHOTORESPONSIVE  DEVICES.  It  is  convenient  to  divide 
photoresponsive  devices  into  two  broad  groups,  namely,  thermal  devices  and  photoelectric 
devices.  In  the  group  of  thermal  devices  are  included  thermojunctions,  bolometers,  and 
radiometers.  In  the  group  of  photoelectric  devices  are  included  photoemissive  cells,  photo- 
conductive  cells,  barrier  photocells,  and  photovoltaic  cells. 

The  thermal  instruments  in  general  respond  to  a  wide  region  of  the  spectrum,  correspond- 
ing to  the  thermal  absorbing  power  of  the  materials  of  which  they  are  constructed  or  with 
which  they  are  coated.  Since  they  are  frequently  coated  with  a  "black"  of  high  and  sub- 
stantially uniform  absorbing  power  through  the  infrared,  visible,  and  ultraviolet  portion 
of  the  spectrum  they  are  often  termed  "non-selective."  They  are  generally  less  sensitive 
than  the  photoelectric  instruments  in  the  spectral  region  to  which  the  latter  respon'd;  they 
are  much  slower  in  response  and,  being  usually  of  relatively  low  resistance,  are  not  easily 
adapted  to  methods  of  electrical  amplification. 

The  'photoelectric  instruments  are  in  general  sensitive  to  relatively  narrow  regions  of  the 
spectrum  (are  strongly  selective),  but  they  are  far  more  sensitive  in  these  regions  than  the 
thermal  instruments.  They  have  also  the  great  practical  advantage  that  they  are  readily 
adaptable  to  the  various  modern  methods  of  electrical  amplification. 


15-02 


THERMAL  DEVICES 


15-03 


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Ther- 


2.  THERMAL  DEVICES 

THERMO  JUNCTIONS.  When  two  different  metals  are  joined  and  the  junction  is 
maintained  at  a  temperature  different  from  the  rest  of  the  circuit,  a  potential  is  generated 
whose  magnitude  is  dependent  on  the  temperature  difference  and  on  the  materials  of  the 
junction.  When  this  difference  is  very  small  the  potential  will  be  proportional  to  the 
temperature.  Bismuth  and  antimony  and  some  of  their  alloys  give  large  potentials  com- 
pared with  most  other  metals,  and  many  thermopiles  have  been  made  of  them.  Good, 
durable  combinations  of  high  sensitivity  commonly  used  at  present 
are  bismuth  against  silver,  bismuth-tin  against  bismuth-antimony 
alloys,  and  Constantin  against  Manganin. 

Other  requirements  beyond  high  thermoelectric  power  are  a  low 
internal  resistance  and  heat  capacity  and  low  thermal  conductivity 
between  the  hot  and  cold  junctions,  and  in  practice  a  compromise 
has  to  be  made  between  these  factors.  Thermoelements  are  used 
in  three  principal  forms:  the  single  junction,  the  multiple-junction 
thermopile,  and  the  radiomicrometer,  the  last  being  a  galvanometer 
with  a  thermojunction  built  directly  into,  and  forming  a  part  of,  the 
moving  coil.  Multiple  junctions  are  preferable  where  the  illumina- 
tion extends  over  some  area;  for  example,  where  spectral  lines  are 
to  be  measured  a  linear  arrangement  of  couples  is  often  used,  a  dia- 
gram of  which  is  shown  in  Fig.  1.  The  warm  junctions,  a,  which 
receive  the  radiation  to  be  measured,  are  commonly  flattened  or 
covered  with  small,  thin  disks  or  squares  of  metal  to  present  a  large 
receiving  area.  These  disks  are  usually  blackened,  for  instance,  with 
a  layer  of  lampblack,  in  order  to  make  them  uniformly  absorptive 
to  a  long  range  of  wavelengths.  The  connection  to  the  cold  junc- 
tions, b  and  &',  must  conduct  as  little  heat  as  possible  and  at  the 
same  time  not  have  too  high  an  electrical  resistance.  In  order  to 
secure  the  maximum  stability  it  is  desirable  to  shield  these  cold 
junctions  thoroughly  from  scattered  radiation  and  to  maintain  their 
temperature  substantially  equal  to  that  of  the  walls  of  the  sur- 
rounding enclosure.  This  is  accomplished  by  attaching  fins  of  relatively  large  area  to 
the  cold  junctions. 

The  galvanometer  should  have  a  resistance  not  far  from  that  of  the  thermopile,  usually 
from  10  to  50  ohms.  There  is  no  advantage  in  increasing  the  number  of  junctions  beyond  a 
certain  point,  and  in  fact  with  a  given  flux  the  overall  sensitivity  may  be  the  largest  with  a 
few  junctions  of  small  heat  capacity  especially  if  they  are  enclosed  in  a  vacuum  which 
eliminates  the  loss  of  heat  by  conduction  and  convection  to  the  atmosphere  and  reduces 
the  disturbing  effect  of  drafts.  The  sensitivity  of  thermopiles  is  expressed  in  volts  per 
unit  of  irradiation,  in  some  cases,  in  terms  of  the  response  from  a  stated  source  at  a  given 

distance,   commonly  a 

Table  1.     Sensitivity  of  Thermopiles  Hefner  candle  at  1  meter. 

Values  for  some  of  the 
standard  thermopiles  are 
given  in  Table  1, 

For  the  sensitivity  in 
watts  per  square  centime- 
ter multiply  the  above  fig- 
ures by  107,  and  for  gram 
calories  per  second  per 
square  centimeter  multi- 
ply by  4.186  X  107-  One 
Hefner  standard  lamp 
with.  14  by  50  mm  diaphragm  opening  gives,  at  1  meter,  9.6  X  10~fi  watt  per  sq  cm  or 
9.6  X  102  ergs  per  sec  per  sq  cm. 

If  a  thermopile  is  to  be  used  with  weak  radiation,  as  it  frequently  is,  it  is  very  important 
that  the  thermopile  and  the  galvanometer  be  adapted  to  each  other  in  order  to  secure  a 
high  sensitivity  of  the  system,  i.e.,  the  ratio  of  the  scale  reading  to  the  flux.  Besides  high 
voltage  sensitivity  of  the  pile  previously  mentioned,  other  i^iiirements  are  that  the 
galvanometer  should  have  a  low  internal  resistance,  high  voltage  sensitivity,  and  a  critical 
damping  resistance  equal  to  that  of  the  pile  inclusive  of  the  leads.  Such  galvanometers 
are  made  by  the  Leeds  and  Northrup  Company,  Philadelphia,  and  Kipp  and  Zonen, 
Holland. 


Ohms 
Resistance 

Volts  per  Erg 
per  Second  per 
Square  Centimeter 

Moll  linear      

20 

0.75  X  10~8 

"FTilger  lin**ftf    .  ,  .  T       -  -               , 

10 

0.7    X  I0~8 

Ivloll  large  surface 

50 

5  0    X  10~s 

Moll  sensitive  vacuum  couple.  .  . 
Moll  quick  vacuum  couple  
Coblentz  vacuum  couple  

45 
20 
14.8 

0.46X  10~8 
0.13  X  I0"s 
0.7    X  NT8 

15-04  ELECTKO-OPTICAL  DEVICES 

The  sensitivity  of  the  system  can  sometimes  be  greatly  increased  by  concentrating  the 
radiation  on  the  junctions  and  also  by  using  one  of  the  numerous  optical  devices  for 
amplifying  the  galvanometer  deflections.  These  are  arrangements  whereby  a  light  beam 
reflected  from  the  galvanometer  mirror  actuates  a  photocell,  the  output  of  which  in  turn 
produces  an  increase  in  the  deflection  or  operates  a  second  galvanometer.  In  the  Moll 
thermorelay  made  by  Kipp  and  Zonen,  or  the  similar  Zernike  differential  couple,  the  re- 
flected light  beam  determines  the  relative  temperatures  of  two  opposing  thermojunctions, 
the  net  output  of  which  controls  a  second  galvanometer.  In  this  manner  the  readings  can 
be  increased  up  to  the  inherent  instability  of  a  galvanometer,  but  care  must  be  taken  to 
preserve  the  linearity  of  the  entire  system. 

BOLOMETER.  The  bolometer  is  essentially  a  sensitive  resistance  thermometer  of  very 
small  heat  capacity;  that  is,  the  electrical  resistance  of  a  fine  wire  or  strip  of  metal  is 
increased  by  the  heating  due  to  the  radiation.  It  was  used  extensively  before  the  develop- 
ment of  the  modern  high-sensitivity,  quick-acting  thermopile,  which  is  more  convenient 
and  stable  under  ordinary  conditions.  In  its  practical  form  two  bolometer  elements  form 
two  arms  of  a  Wheatstone  bridge,  which  the  radiation  unbalances.  It  is  essential  that  a 
material  such  as  platinum  be  used  which  has  a  high  temperature  coefficient  of  resistance 
as  well  as  a  low  heat  capacity  and  conductivity.  Unless  unusual  precautions  are  taken 
the  entire  bridge  network  is  subject  to  temperature  fluctuations  which  render  the  readings 
uncertain.  The  sensitivity  of  bolometers  is  about  one-millionth  of  a  degree  per  millimeter 
deflection  of  a  galvanometer  used  without  intermediate  amplification.  For  example,  a 
bolometer  of  2,8  ohms  resistance  and  using  40  mils  current  gave  a  deflection  of  45  cm 
when  exposed  to  1  candle  at  1  meter  with  a  galvanometer  sensitivity  of  1.5  X  10~10  amp 
per  mm,  the  scale  being  at  1  meter. 

THERMISTOR  BOLOMETER.  A  more  recently  developed  form  of  bolometer  is  one 
made  of  thermistor  materials,  that  is,  semiconductors  whose  resistance  varies  rapidly 
with  temperature.  Combinations  of  oxides  of  nickel,  manganese,  and  cobalt  change  their 
resistance  about  4  per  cent  per  degree  centigrade,  or  about  ten  times  as  much  as  platinum. 
The  oxides  are  prepared  in  the  form  of  thin  flakes  cemented  to  glass  or  quartz.  A  typical 
flake  3  mm  long,  0.2  mm  wide,  and  0.01  mm  thick  has  a  resistance  of  4  X  106  ohms  and 
with  250  volts  applied  gives  a  sensitivity  of  300  volts  per  incident  watt  or  18  volts  per 
watt  per  sq  cm.  The  spectral  response  is  determined  by  the  optical  properties  of  the  oxide 
constituents,  which  may  show  regions  of  relative  transparency  in  the  infrared.  Because 
of  the  high  resistance  the  output  of  the  thermistor  bolometer  is  well  adapted  to  electrical 
methods  of  amplification. 

RADIOMETER.  The  Nichols  radiometer  is  a  self-contained  instrument  consisting  of 
two  similar  vanes  of  blackened  mica  or  platinum  on  a  horizontal  arm  and  suspended  in  a 
vacuum.  It  is  a  development  of  the  toy  known  as  Crookes'  radiometer  frequently  seen 
in  optical  shops,  which  consists  of  four  vanes,  each  blackened  on  one  face,  on  arms  balanced 
on  a  needle-point,  which  rotate  when  illuminated.  The  behavior  of  the  radiometer  is 
dependent  on  the  gas  pressure;  at  higher  values  the  blackened  sides  of  the  vanes  are  drawn 
in  turn  toward  the  window;  at  lower  pressures  the  warming  of  the  blackened  faces  by  the 
radiation  causes  the  residual  gas  molecules  to  rebound  from  their  surface  directly  to  the 
cooler  window  and  push  the  vanes  away  from  it  and  the  radiating  source.  All  practical 
instruments  have  the  gas  pressure  so  adjusted  that  they  work  by  the  latter  method.  The 
Nichols  radiometer  is  used  by  measuring  the  deflection  of  the  vanes  by  means  of  a  small 
mirror  attached  to  the  cross-arm  supporting^  them,  its  rotation  being  observed  by  the 
usual  telescope  and  scale.  The  sensitivity  of  the  radiometer  to  radiation  is  of  the  same 
magnitude  as  that  of  the  bolometer  with  a  sensitive  galvanometer, 

SPEED  OF  RESPONSE  OF  THERMAL  DEVICES.  The  thermal  devices  are  in 
general  slow  to  respond  to  variations  in  signal  strength  and  hence  are  not  well  adapted  for 
following  rapidly  fluctuating  radiation,  such  as  radiation  modulated  at  speech  frequencies. 
The  time  constant,  defined  as  the  interval  in  which  the  response  declines  to  1/e  value,  is 
for  the  fastest  devices  of  the  order  or  5  milliseconds. 


3.  PHOTOEMISSIVE  CELLS 

STRUCTURE.  In  photoemissive  cells  an  electropositive  metal  surface  is  placed  in  a 
highly  evacuated  enclosure,  usually  of  glass  or  quartz,  together  with  another  metal  plate, 
the  electropositive  material  constituting  the  sensitive  cathode,  the  other  plate  the  anode, 
and  both  plates  are  connected  with  terminals  led  through  the  glass.  The  action  of  the 
cell  is  as  follows:  When  light  falls  on  the  cathode,  electrons  are  emitted  into  the  space 
above.  These  pass  over  to  the  anode,  and,  if  the  terminals  outside  the  cell  are  connected 
through  a  current-measuring  device,  a  current  is  observed  which  varies  with  the  total 


PHOTOEMISSIYE  CELLS 


15-05 


illumination.  Although  some  current  will  flow  without  a  battery  being  connected  in  series, 
it  is  common  practice  to  use  one,  and  then  the  cells  act  primarily  as  valves,  tlie  illumination 
controlling  the  amount  of  current  which  is  permitted  to  pass. 

Practically  all  photoemissive  cells  use  alkali  or  alkaline-earth  metals  as  their  light-sensi- 
tive materials,  and  the  structure  is  largely  influenced  by  the  problem  of  introducing  these 
metals  into  the  glass  or  quartz  enclosure.  In  early  types  of  photoemissive  cells  an  alkali 
metal,  such  as  sodium  or  potassium,  was  introduced  in  molten  form  into  a  simple  spherical 
bulb  provided  with  one  leading-in  wire  in  contact  with  the  pool  of  alkali  metal^and  a 


to 
Central  Cathode  Cell 


Central  Anode  Cells 


Jfehf 


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selenium  Conductive 
Cell 


CO 
Voltaic  Cell 

FIG.  2.    Types  of  Photo-responsive  Cells 


Photomultiplier  Tube 


second  wire  to  serve  as  an  anode.  This  type  has  been  almost  entirely  superseded  by  cells 
in  which  the  alkali  metal  forms  a  thin  film  upon  some  other  metal.  Two  general  types 
of  structure  are  common,  respectively,  the  central-anode  and  the  central-cathode  types. 
The  central-anode  type  is  represented  by  cells  (a)  and  (&),  Fig.  2.  In  these  the  alkali 
metal  has  been  distilled  upon  a  metal  plate  or  on  the  walls  of  a  bulb  with  an  opening  left 
for  a  window  for  the  light.  The  central  anode  consists  of  a  wire  loop  or  grid.  The  central- 
cathode  type,  which  is  illustrated  in  (c),  Fig.  2,  consists  of  a  metal  plate,  on  which  the  alkali 
metal  has  been  deposited,  more  or  less  surrounded  by  a  grid  or  metal  covering  on  the  bulb 
wall  serving  as  the  anode. 

ELECTRODE  MATERIALS.  The  most  commonly  used  photosensitive  materials  are 
the  alkali  metals,  sodium,  potassium,  rubidium,  and  cesium,  whose  intrinsic  sensitiveness 
increases  in  the  order  given.  The  sensitiveness  of  the  pure  metals  is,  however,  far  below 
that  of  the  same  metals  when  given  various  special  treatments,  such  as  being  exposed  to  a 
glow  discharge  in  hydrogen  or  being  put  down  upon  an  oxidized  base  and  given  special 


15-06 


ELECTRO-OPTICAL  DEVICES 


54321 

-f-  Volfs  on  Cafhode 

FIG.  3. 


1234 
—  Volts  on  Cathode 


heat  treatments.  The  most  generally  used  cell  at  present  consists  of  a  silver  plate  which  is 
oxidized  and  subsequently  exposed  to  cesium  vapor  and  heat  treated.  A  later  develop- 
ment has  been  the  combination  of  antimony  with  cesium,  which  produces  a  high  sensitivity 
localized  in  the  blue  region  of  the  spectrum.  Other  materials,  such  as  barium  and  cadmium, 
are  occasionally  used  when  sensitiveness  to  particular  regions  of  the  spectrum  is  required. 
VACUUM  AND  GAS-FILLED  CELLS.  Photoemissive  cells  are  commonly  made  up 
either  as  high-vacuum  cells  or  as  gas-filled  cells.  In  high-vacuum  cells,  the  photoelectric 

current  consists  entirely  of  the  elec- 
trons emitted  from  the  cathode.     In 
gas-filled  cells,  an  inert  gas  such  as  ar- 
gon is  introduced  at  a  pressure  of  sev- 
eral tenths  of  a  millimeter  of  mercury, 
-^      and,  when  a  sufficient  voltage  is  ap- 
plied, the  photoelectrons  by  collision 
with     the     gas    produce    ionization, 
whereby  the  current  is  amplified  sev- 
eral times.    This  process  of  gas  ampli- 
fication  is  limited  by  the  ignition  volt- 
age, at  which  a  visible  glow  discharge 
takes  place,  which  is  self-perpetuating 
Current-voltage  Relation  for  a  Central  Cathode   an(}    ^^   injure    the    surface    of   the 

CeU  cathode. 

VOLTAGE-CURRENT  CHARACTERISTICS.  The  relationship  between  voltage  and 
photoelectric  current  under  constant  illumination  depends  upon  the  physical  structure  of 
the  cell,  particularly  upon  the  size  and  arrangement  of  the  electrodes.  The  structure 
best  adapted  for  studying  the  fundamental  phenomena  of  photoelectricity  is  the  central- 
cathode  arrangement.  The  voltage-current  relation  for  a  vacuum  central-cathode  cell, 
in  which  the  dimensions  of  the  cathode  are  negligibly  small  compared  with  those  of  the 
anode  which  encloses  it  as  completely  as  is  possible  while  allowing  space  for  the  entrance 
of  light  and  mechanical  parts,  is  shown  in  Fig.  3.  In  this  figure,  the  abscissas  represent 
voltages  applied  to  the  cathode  and  show  that,  at 
a  certain  positive  voltage,  that  is,  with  a  field  which 
opposes  the  emission  of  electrons,  the  photoelectric 
current  makes  its  appearance.  This  point,  which 
is  a  measure  of  the  maximum,  energy  given  to  the 
photoelectrons  by  the  incident  light,  is  called  the 
"stopping  potential.*'  As  the  positive  voltage  is 
decreased,  the  photoelectric  current  increases  until 
it  reaches  a  steady  value  at  the  saturation  voltage. 
This  voltage  will  be  zero  if  anode  and  cathode  are 
of  the  same  material,  but  it  will  be  displaced  by  the 
contact  difference  of  potential  between  the  anode 
and  cathode  where  the  materials  are  different, 

In  Fig.  4  is  shown  the  corresponding  character- 
istic for  a  vacuum  central-anode  cell.  Here,  be- 
cause of  the  small  target  presented  by  the  anode, 
saturation  is  reached  only  at  high  voltages.  In 
Fig.  4  is  also  shown  the  voltage-current  relation- 
ship for  a  gas-filled  cell  of  the  same  construction. 
At  low  voltages  this  characteristic  is  essentially 
that  determined  by  the  structure  of  the  cell, 
whether  it  be  central  anode  or  central  cathode, 
but  at  higher  voltages  the  current  increases  rapidly 
up  to  the  ignition  voltage  of  the  gas  which  is,  in 
general,  of  the  order  of  magnitude  of  several  hun- 
dred volts.  Beyond  this  point,  a  sustained  dis- 
charge occurs  with  a  negative  voltage-current 
characteristic. 

ILLUMINATION-CURRENT  RELATIONSHIP.  With  an  ideal  cell  structure,  the 
number  of  electrons  released  by  the  light,  and  consequently  the  photoelectric  current,  are 
directly  proportional  to  the  illumination.  In  all  practical  cells,  however,  this  strict  rela- 
tionship is  departed  from  to  a  slight  extent  because  of  charging  effects  of  exposed  glass 
walls  and  other  obscure  phenomena.  For  this  reason,  photoelectric  cells  are  applicable  to 
precision  photometric  measurement  only  if  their  exact  characteristics  are  determined  by 
experiment,  or  if  a  substitution  method  is  used.  In  gas-filled  cells,  the  illumination-current 


20 


18 


16 


20        40 


s_ 


7 


60 
Votts 


80       100      120 


FIG.    4.     Current-voltage    Relation    for 
Gas  and  Vacuum  Central  Anode  Cells 


PHOTOEMISSIVE  CELLS 


15-07 


relationship  departs  from  strict  proportionality  whenever,  as  is  common,  a  high  series 
resistance  is  used  either  as  part  of  the  measuring  system  or  for  protection  against  the 
occurrence  of  a  glow  discharge.  The  photoelectric  current  flowing  through  this  series 
resistance  uses  up  part  of  the  applied  potential,  thereby  lowering  the  voltage  across  the 
cell  itself,  which  accordingly  works  upon  a 
lower  point  in  the  voltage-current  character- 
istic of  Fig.  4.  Typical  illumination-current 
curves  for  a  gas-filled  cell  with  various  series 
resistances  are  shown  in  Pig.  5. 

WAVELENGTH  RESPONSE.  The  photo- 
electric current  per  unit  of  incident  radiation 
varies  greatly  with  wavelength  and  in  different 
ways,  depending  on  the  characteristics  of  the 
sensitive  material  employed. 

Figure  6  shows  the  equi-energy  response 
curves  for  several  typical  cells  compared  with 
the  average  eye.  These  exhibit  maxima  of 
emission  strongly  localized  in  different  parts  of 
the  spectrum.  The  maximum  lies  in  the  blue 
for  the  potassium  hydride  cell,  in  the  infrared 
for  the  cesium-silver  oxide  cell,  and  in  the  near 
ultraviolet  for  the  cesium-antimony  cell.  The 
type  of  cell  to  choose  for  a  given  purpose  to 
secure  the  maximum  response  depends  on  the 
characteristics  of  the  light  source  used.  Sources 
like  the  tungsten  lamp,  whose  emission  is 
greatest  in  the  infrared,  evoke  a  maximum 
response  from  cells  of  the  cesium  oxide  type, 
as  illustrated  in  Fig.  7,  where  two  cells  of  Fig.  6 
are  excited  by  tungsten  lamplight  instead 
of  an  equal-energy  spectrum.  Daylight  and  the  quartz  mercury  arc,  on  the  other  hand, 
evoke  a  greater  response  from  cells  whose  sensitiveness  is  farther  toward  the  blue  end  of 
the  spectrum. 

SENSITIVITY.    The  luminous  sensitivity  of  photoemissive  cells  is  ordinarily  defined 
by  their  output  in  microamperes  per  lumen  of  steady  light.   For  gas  cells  which  have  some 


Fia.    5.     Currenfc-iUnTnmatlon    Relation    for 
Gas  Cell  with  Various  Series  Resistances 


Relative  P.E.  current  per  unit  energy,  maximum  =1,0 
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00           4000           5000           6000           7000           8000           9000         10,000        11,000        12,000 

Wavelength,  angstmm  units 

PIG,  6.     Equi-energy  Spectral  Response  for  Some  Types  of  Blue-  and  Red-sensitive  Cells  andjiihe 

Average  Eye 

inertia  at  acoustic  frequencies  this  sensitivity  is  often  stated  in  terms  of  the  cell's  ability 

to  follow  a  sinusoidally  varying  light  at  one  or  more  stated  frequencies  within  this  range. 

In  general  the  sensitivity  depends  on  the  cell  voltage,  the  flux  intensity  and  its  color,  the 

resistance  in  the  external  circuit,  and  the  distribution  of  the  flux  over  the  cathode  if  it  is 


15-08 


ELECTRO-OPTICAL  DEVICES 


20 


18 


16 


-12 
£ 

010 


§  8 


CsO 


non-uniform,  as  it  frequently  is  to  some  extent.  No  allowance  is  ordinarily  made  for  the 
glass  bulb  or  any  absorbing  films  on  its  interior  as  they  must  be  tolerated  in  using 
the  cell. 

Since  the  output  is  dependent  on  the  spectral  composition  of  the  light,  this  should  be 
specified;  as  tungsten  lamps  are  ordinarily  used  in  rating  cells,  it  is  done  by  stating  their 

color  temperature.  There  is  no 
general  agreement  on  a  standard 
color  temperature,  but  287 0  deg 
abs  has  been  suggested  and  is  used 
by  some  manufacturers.  Figure  8 
shows  how  the  sensitivities  of  the 
red-responsive  cesium  oxide  and 
the  blue-responsive  potassium  hy- 
dride cells  vary  with  this  tempera- 
ture. The  sensitivity  should  not 
be  confused  with  the  total  output 
obtainable  from  a  cell  as  the  fila- 
ment temperature  of  the  lamp  is 
raised;  this  output  always  increases 
rapidly.  The  current  per  lumen  of 
light,  however,  decreases  with 
highly  red-sensitive  cells  because 
there  is  relatively  less  red  to  blue 
or  total  light.  Typical  sensitivities 
are  given  in  Table  2. 

The  sensitivity  of  photocells  is 
also  defined  by  the  magnitude  of 
their  response  per  unit  radiant  flux 

_  _        at  a  given  wavelength.    For  exam- 

3000  4000  5000  6000  70OO  sooo  9000  loooo  uooo  12000  pie,  the  cells  in  Fig.  6  may  be  so 
Wave-length  -  A°  evaluated    by   their   microampere 

FIG.  7.    Tungsten  Spectral  Response  for  KTT  and  CsO  Cells    output  per  microwatt  of  irradiation 

at  the  wavelengths  of  maximum 

response.  Typical  values  for  xinamplified  commercial  cathodes  are:  SI,  0.002  at  7500  A; 
S2,  0.002  at  8000  1;  83,0.002  at  4400  A;  S4,  0.04  at  3750  i. 

In  evaluating  photoemissive  cells  the  permanence  and  stability  of  the  sensitivity  must 
be  considered  along  with  its  absolute  value.  Commercial  emissive  cells  are  usually  per- 
manent and  stable  provided  that  excessive  voltages  are  not  used  (particularly  on  gas  cells) 


c 
o 

I 

\ 

\ 

Sensitivity  In  Microamperes  por 

5  8 

\ 

^ 

^ 

"tX^Cej 

ium 

^-^ 

^^ 

^r— 

^ 

^^ 

Potass 

ium 

2000°  4000°  6000°  8000°  Abs. 

Temperature  of  the  Light  Source 
Fia.  8.     Variation  of  Sensitivity  with  Color  Temperature 

and  provided  that  they  are  not  subjected  to  excessive  heating  or  too  concentrated  illumina- 
tion. Any  ionization  even  of  residual  vapor  or  gas  must  be  avoided  for  maximum  stability 
ao  that  in  precision  use  anode  voltages  as  low  as  20  volts  are  recommended. 

e,  due  to  a  conducting  film  around  the  stem  or  to  thermionic  emission  from  the 


PHOTOBMISSIVE   CELLS 


15-09 


cathode,  may  render  high  sensitivity  more  or  less  useless.     Cathodes  deficient  in  red 
sensitivity,  such  as  cesium-antimony,  have  greatly  reduced  thermionic  emission. 

FREQUENCY  RESPONSE.  The  emission  of  electrons  in  the  photoelectric  effect  is 
practically  instantaneous,  and  accordingly  the  emission  should  be  capable  of  following 
intermittent  illumination 

Table  2.     Sensitivity  of  Photoemissive  Materials 
(Tungsten  lamp  source) 


up  to  exceedingly  high  fre- 
quencies, such  for  instance 
as  those  required  for  tele- 
vision. In  practice,  the 
frequency  response  drops 
off  at  high  frequencies, 
owing  either  to  the  pres- 
ence of  the  gas  in  gas-filled 
cells  or  in  vacuum  cells  to 
the  presence  of  the  high 
series  resistance,  which  is 
ordinarily  used  for  cou- 
pling purposes,  or  to  the 
electrostatic  capacitance 
of  the  cell  considered  as  a 
condenser.  Typical  fre- 
quency-response curves 
are  given  in  Fig.  9. 


Microamperes 
per  Lumen  * 

Filament  Color 
Temperature, 
deg  Kelvin 

Potassium  hvdride 

0  5-1   0 

2848 

Potassium-sulfur    . 

I  8 

2848 

Sodium-sulfur  

">  4 

2848          j 

Sodium-sulfur-oxygen    .  .          ... 

6-8 

2848           i 

Cesium  antimony  (34)  

50-45 

2870 

Cesium  oxide  on.  silver  (S  1  ,  S2)    ... 
S3     ... 

10-50 
6  5 

2870 
2870 

S5  .  . 

15  0 

2870 

S8  

3.0 

2870 

*  Intrinsic  sensitivity  of  the  cathode;  amplification  by  gas  or  by  sec- 
ondary emission  may  increase  these  figures  by  several  times  or  by  sev- 
eral orders  respectively. 


In  a  gas  cell  the  frequency  response  is  dependent  on  the  applied  voltage,  and  if  this  is 
near  the  breakdown  the  loss  of  response  may  considerably  exceed  the  values  shown.  It 
is  not  advisable  to  exceed  the  voltage  recommended  by  the  manufacturer. 

MEASURING  CIRCUITS  FOR  USE  WITH  PHOTOEMISSIVE  CELLS.  There  are 
two  general  methods  of  measuring  photoelectric-cell  output:  first,  the  measurement  of 
the  current  directly;  and  second,  the  measurement  of  the  voltage  drop  across  a  series 
resistance.  The  photoelectric  current  is  measured  by  inserting  a  sensitive  galvanometer 
in  series  with  the  cell  and  a  battery,  and  the  method  is  limited  in  sensitiveness  only  by  the 
sensitivity  of  available  galvanometers.  The  voltage  drop  across  a  high  resistance  is 
measured  by  means  of  an  electrometer  or  of  special  vacuum  tubes  designed  to  function  in 
the  same  manner.  For  extremely  minute  illuminations,  the  high  resistance  may  be  made 
infinite,  and  the  current  may  be  ascertained  by  the  rate  at  which  the  electrometer  or 
equivalent  device  charges  up. 

A  number  of  d-c  galvanometers  are  made  which  are  suitable  for  use  with  photoemissive 
cells,  ranging  in  sensitivity  down  from  about  10~10  amp;  this  can  be  extended  to  about 
10 ~12  amp  with  a  device  like  the  Moll  thermorelay.  The  resistance  of  the  galvanometer 
is  immaterial,  provided  that  it  has  a  high  current  sensitivity,  because  the  resistance  of  the 
cell  will  ordinarily  be  enormously  greater.  It  is  not  desirable  for  several  reasons  to  use 

an  instrument  that  is  much 
more  sensitive  than  the  work 
demands,  but  the  testing  of 
various  types  of  cells  of  widely 
varying  characteristics  with 
different  colors  and  intensities 
of  iHurnination  may  require  a 
very  flexible  galvanometer 
system.  In  this  event,  shunts 
can  be  used  if  they  are  so  ar- 
ranged as  not  to  interfere  with 
the  critical  damping.  A  good 
method  is  to  insert  various 
sections  of  the  damping  resist- 
ance in  the  cell  circuit.  The 
Leeds  and  Northrup  type 
22S5-F  galvanometer  is  suit- 
able for  this  purpose.  Where 
considerable  current  is  available  and  portability  is  desired,  microammeters  of  the  Rawson 
or  Weston  type  are  convenient,  especially  when  provided  with  a  dial  or  other  means  of 
securing  various  scale  sensitivities. 

In  all  cases,  great  care  must  be  taken  to  protect  any  instrument  from  a  breakdown  of 
the  cell  by  inserting  sufficient  series  resistance,  at  the  same  time,  if  possible,  retaining 
linearity  of  response.  For  work  of  the  highest  sensitivity  it  is  necessary  to  resort  to  an 


LOO 

% 

50 

20 

0 
2 

—  r~  ^^ 

> 

Gas  Cell 
Vacuum  Cell 

DO                    506             1000           20OQ                  5000          1C,OOO 

FIG.  9. 


Frequency 
Frequency  Response  for  Gas  and  Vacuum  Cells 


15-10  ELECTRO-OPTICAL  DEVICES 

electrometer  such  as  the  Compton  type,  the  applied  potential  being  secured  by  the  drop 
across  several  megohms  in  series  with  the  cell,  or  by  using  the  cell  as  a  constant  current 
source  and  measuring  the  rate  of  charging  of  the  electrometer  placed  in  series.  Effective 
use  of  the  electrometer  requires  a  permanent  laboratory  installation.  Where  greater 
portability  is  desired  practically  equivalent  results  can  be  secured  by  amplifying  the  very 
small  currents  by  an  "FP.54  pliotron"  and  measurement  on  a  moving-coil  galvanometer 
such  as  the  Leeds  and  Northrup  type  R.  For  the  methods  of  using  an  electrometer  see 
Hughes  and  DuBridge's  Photoelectric  Phenomena,  1932,  pp.  435-444,  and  for  the  corre- 
sponding technique  of  amplification  see  DuBridge,  Physical  Review,  Vol.  37,  Feb.  15,  1931, 
pp.  392-400.  For  descriptions  of  the  more  common  electrometers  see  the  catalog  of  the 
Cambridge  Instrument  Company,  and  for  galvanometers  consult  catalogs  of  Leeds  and 
Northrup  Company  and  Kipp  and  Zonen. 

AMPLIFICATION  OF  PHOTOEMISSIVE-CELL  OUTPUT.  Photomultiplier  Tube. 
The  initial  current  produced  by  illumination  of  a  photocathode  may  be  greatly  amplified 
by  utilizing  the  phenomenon  of  secondary  emission.  In  the  photomultiplier  tube  the  elec- 
trons emitted  from  the  cathode  are  directed  by  a  suitable  high-voltage  field  onto  another, 
usually  similar,  electrode  where  they  cause  the  emission  of  secondary  electrons  which  are 
greater  in  number  than  the  impinging  electrons.  This  process  may  be  repeated  a  number 
of  times.  Present  commercial  multiplier  tubes  have  as  many  as  nine  such  stages,  and 
even  larger  numbers  have  been  used  in  special  tubes.  The  overall  current  amplification 
thus  produced  is  of  the  order  of  several  hundred  thousand  times.  A  typical  electron  mul- 
tiplier structure  is  shown  in  (g),  Fig.  2t  and  commercially  available  models  are  listed  in 
Table  3,  p.  15-16. 

Circuits  for  Amplifying  Photoemissive-cell  Output.  The  photoemissive  cell,  because 
of  its  exceedingly  high  internal  resistance,  is  admirably  adapted  for  use  in  connection  with 
vacuum-tube  amplifying  devices.  In  the  simplest  arrangement,  the  electrometer  described 
in  the  previous  section  is  replaced  by  the  grid  of  a  three-electrode  tube,  and  the  potential 
acquired  by  it  modulates  the  current  through  the  vacuum  tube,  which  may  be  further 
amplified  by  successive  stages. 

Circuits  for  amplifying  photoerm aai ve-cell  output  are  determined  by  the  type  of  appli- 
cation of  the  cell.  These  can  be  put  largely  into  three  classes:  trigger  operation,  d-c  linear 
operation,  and  a-e  linear  operation.  The  first  class  is  concerned  with  merely  a  qualitative 
response,  and  there  is  no  particular  requirement  for  linearity.  A  large  number  of  uses* 
come  under  this  classification,  such  as  the  operation  of  relays  in  various  counting  and  sort- 
ing processes.  The  usual  circuit  for  amplification  consists  in  feeding  the  voltage  drop 
across  the  cell  load  into  a  thyratron  tube  which  in  turn  actuates  a  relay  (see  Section  21). 
The  necessity  for  supplementary  amplification  will  depend  on  the  light  variation  available 
and  the  marginal  requirements  of  the  relay.  It  is  necessary  to  provide  for  the  release  of 
the  thyratron,  and  this  can  be  easily  done  by  using  alternating  current  on  its  cathode  or 
interrupting  the  direct  current.  The  sensitivity  of  trigger  systems  can  ordinarily  be  in- 
creased by  the  use  of  large  load  resistances  in  the  cell  circuit. 

A  method  of  use  closely  allied  to  the  preceding  in  the  characteristics  demanded  of  the 
cell  is  as  a  null  device  in  substitution  photometry,  where  the  only  requirement  is  a  suitable 
recorder  with  enough  amplification  to  give  the  necessary  precision.  If  the  test  and  com- 
parison lights  are  rapidly  alternated  on  the  cell,  the  method  permits  of  a-c  amplification 
with  its  attendant  advantages  of  great  efficiency  and  simplicity,  the  match  being  given 
by  zero  a-c  output. 

The  second  class,  d-c  linear  operation,  covers  the  direct-reading  method  of  photometry, 
an  example  of  which  is  the  recording  of  daylight  intensity.  This  method  has  the  disad- 
vantage of  requiring  stable  cells  of  reproducible,  linear  characteristics,  a  requirement  not 
always  easy  to  meet  for  precision  photometry.  For  many  purposes,  however,  the  require- 
ments are  not  rigid  and  commercial  cells  are  suitable  for  the  purpose.  Several  methods 
of  amplification  are  possible.  One  is  to  use  straight  d-c  resistance-coupled  amplification, 
preferably  at  low  cell  currents,  with  an  electrometer  tube  as  the  first  stage.  In  order  to 
minimise  the  tendency  to  instability  inherent  in  d-c  amplification,  use  is  sometimes  made 
of  balanced  d-c  amplification.  At  low  light  intensities  these  direct  methods  require  much 
care  to  guard  against  leakages  in  and  around  the  cell,  and,  for  some  purposes,  specially 
constructed  cells  are  necessary.  Another  precaution  that  must  be  taken  is  to  insure  the 
linearity  of  response  of  the  amplifier.  A  third  method  of  amplification  that  is  sometimes 
applicable  is  to  interrupt  the  illumination  of  the  cell  and  use  a-c  amplification  (see  Section 
7).  This  has  the  advantages  of  greater  stability,  of  minimizing  leakages,  and  of  securing 
the  efficiency  of  interstage  coupling  by  transformers. 

Such  uses  of  photocells  as  for  picture  transmission  and  sound  pictures  may  be  classed 
as  a-c  linear  operation.  Here  the  response  must  be  not  only  linear  but  also  uniform  over  a 
range  of  frequencies  which  may,  as  in  television,  be  very  large.  This  imposes  certain 


PHOTOCONDUCTIVE   CELLS 


15-11 


restrictions  on  the  load  impedance  in  the  cell  circuit.  Since  a  large  value  is  necessary  in 
order  to  secure  a  high  voltage  output  into  the  amplifier,  the  shunting  capacitance  becomes 
very  important  and  severely  limits  the  useful  value  of  the  impedance  that  can  be  used- 
Noise  Limit  to  the  Use  of  Amplification.  Broadly  speaking,  any  photoelectric  current, 
however  minute,  can  by  successive  amplification  be  raised  to  any  desired  high  value.  A 
limit  is  set  to  the  effectiveness  of  this  process  by  the  noise  in  the  cell  or  its  associated 
circuits,  which  is  amplified  along  with  the  signal.  The  significant  specification  of  sensitivity 
of  a  photocell  thus  becomes  the  signal  which  can  override  the  noise.  Noise  in  a  photocell 
exists  because  of  natural  fluctuations  of  current  at  low  values  and  by  the  thermal  agitation 
of  electrons  in  the  coupled  resistances,  and  it  is  a  function  of  temperature,  frequency,  and 
band  width.  In  addition  there  are  many  extraneous  sources  of  noise  from  such  causes 
as  the  ionization  in  gas  tubes,  interference,  microphonic  contacts,  power  supplies,  and 
dielectric  leakages.  These  can  usually  be  diminished  to  secondary  importance  by  careful 
shielding  and  design  of  the  circuits.  Where  considerable  amplification  of  weak  photo- 
currents  of  wide  bandwidth  is  required  there  will  be  a  gain  hi  the  signal-to-noise  ratio  by 
using  a  multiplier  phototube  for  the  initial  stages. 


4.  PHOTOCONDUCTIVE  CELLS 

The  fact  that  light  could  directly  change  the  electrical  resistance  of  a  substance  was 
first  discovered  about  1880  by  observation  of  the  effect  in  metallic  selenium,  and  since 
that  time  some  2000  papers  have  been  published  concerning  its  behavior  and  use.  Never- 
theless, the  mechanism  whereby  light  releases  electrons  remains  obscure.  Furthermore, 
many  of  its  characteristics  depend  to  a  considerable  extent  on  the  method  of  const  rucrion 
as  well  as  the  conditions  under  which  they  are  measured.  Two  types  of  construction  have 
been  used  for  such  cells:  one  in  which  a  thin  layer  of  selenium  is  sandwiched  between  two 
electrodes,  one  of  which  must  be  translucent  to  permit  illumination  of  the  layer;  and  one 
in  which  two  interlocking  metallic  grids  or  combs  are  bridged  by  a  layer  of  selenium.  Mod- 
ern cells  are  usually  of  this  second  type,  although  the  barrier  cells  described  below  use  the 
first.  An  example  of  a  conductive  selenium  cell  of  comb  construction  is  shown  in  Fig.  2(d  )  . 
Other  materials  which  exhibit  similar  properties  are  thallium  sulfide  and  lead  sulfide.  The 
former  is  used  in  a  cell  known  commercially  under  the  name  of  "Thalofide."  Both  these 
newer  cells  possess  properties  which  render  them  superior  to  selenium. 

Current-illumination  Relationship.  If  a  selenium  cell  is  placed  in  series  with  a  battery 
and  meter  and  is  illuminated  with  increasing  intensity,  the  resulting  current  will  usually 
be  of  the  shape  shown  in  Fig.  10.  In  general 
the  change  in  conductance,  G,  follows  the 
equation 


where  g  and  x  are  constants,  F  the  light  flux, 
TQ  the  dark  resistance,  and  r»-  the  light  resist- 
ance. The  constant  x  is  frequently  about 
0.5,  so  that  the  photocurrent  varies  approxi- 
mately as  the  square  root  of  the  illumina- 
tion. The  dark  resistances  of  different  grid 
cells  vary  greatly  but  usually  are  from 
100,000  up  to  several  megohms;  those  of  the 
sandwich  type  are  much  lower  and  may  be 
only  a  few  hundred  or  thousand  ohms.  The 
current-illumination  relation  of  thallium 
sulfide  differs  from  selenium  in  not  being 
curved  as  strongly  toward  the  illumination 
axis. 

Photoconductive  cells  are  made  to  oper- 
ate on  a  variety  of  applied  voltages,  and  in 
general  the  recommendations  of  the  maker 
should  be  followed.  If  not  hermetically  sealed  the  cells  should  be  protected  from  excessive 
moisture  and  corrosive  gases  such  as  sulfur  fumes.  High  illumination  also  causes  deterio- 
ration of  some  cells. 

WAVELENGTH  RESPONSE.  Typical  curves  of  the  spectral  response  for  both  selenium 
and  thallium  sulfide  are  shown  in  Fig.  11.  It  is  characteristic  of  the  former  to  have  a 
peak  at  7000  or  7500  angstroms,  the  sensitivity  up  through  the  visible  region  being  variable 


1 

s 

o 

( 
F* 

^ 

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^ 

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J^ 

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s 

I/ 

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j 

312345678 
Illumination 

3.    10.     Current-Illumination  Response  for  Se- 

lenium  Conductive  Cell 


15-12 


ELECTRO-OPTICAL  DEVICES 


with  different  cells  and  frequently  rising  again  as  the  ultraviolet  is  approached.  Thallium 
sulfide  is  much  more  infrared-sensitive  than  selenium,  with  a  maximum  around  10,000 
angstroms,  after  which  it  falls  off  rapidly,  whereas  lead  sulfide  has  a  maximum  at  25,000 

angstroms  and  falls  to  20  per  cent  at  4000  and 
33,000  angstroms. 

FREQUENCY  RESPONSE.  Many  oscil- 
lographic  observations  have  been  made  on 
the  speed  with  which  the  photocurrent  builds 
up  when  a  selenium  cell  is  suddenly  'illumi- 
nated and  on  the  rate  of  decay  of  the  current 
when  the  light  is  removed.  This  method  of 
observing  its  inertia,  however,  is  ordinarily 
not  so  useful  as  the  method  of  measuring  its 
response  to  continuously  interrupted  light  of 
known  frequencies.  Figure  12  shows  such  a 
measurement  starting  at  very  low  and  extend- 
ing to  nearly  10,000  interruptions  per  second, 
the  ordinates  being  the  a-c  response  relative 
to  the  flat  portion  as  unity.  The  most  re- 
cently developed  Thalofide  cells  fall  off  much 
more  slowly  with  frequency  and  are  useful  up 
to  the  lower  voice  frequencies;  the  frequency 
performance  of  lead  sulfide  is  notably  better  than  that  of  thallium  sulfide,  remaining 
practically  constant  up  to  5000  cps. 

SENSITIVITY.  Photoconductive  cells  are  commonly  rated  by  then*  ratio  of  dark-to- 
light  resistance  at  a  stated  iUurnination  and  voltage.  Thus  a  cell  may  be  stated  to  have  a 
ratio  of  6  at  100  ft-c  using  a  bat- 
tery of  100  volts.  As  there  is 
no  agreement  on  a  standard 
light  intensity  or  color,  care 
must  be  exercised  in  comparing 
cells  from  different  manufac- 
turers. The  d-c  sensitivities 
may  be  divided  into  two  classes 
according  to  whether  the  load 
in  series  with  the  cell  is  a  relay 
to  be  operated  directly,  or  a 
resistance,  the  voltage  drop 
across  which  is  in  turn  required 


40OO   50OO  6000   7000  SOOO  9OOO  10000  11000 
Wave-length 

FIG.  11.     Spectral  Response  for  Selenium  and 
Thallium  Sulfide  Conductive  Cells 


0.1 


0.01 


0.001 


i 


i 


1 


T 


_]_ 


I 


'  1 


OJ. 


102          103 


to  operate  the  grid  of  a  therm- 
ionic tube.    In  the  latter  case 


i          10 

Frequency 
FIG.  12.    Frequency  Loss  for  Selenium  Conductive  .Cell 


the  expression  for  the  voltage  sensitivity,  a;  of  selenium  is 
dEe       Eng  __ 


where  Es  —  voltage  across  load  effective  on  grid. 

E  =  battery  voltage. 

F  =  light  flux. 

TO  =  dark  resistance  of  cell.  

g  =  constant  of  the  cell,  its  light  conductance  being  g\/F. 

Ti  =  light  resistance  of  cell. 

This  is  the  equation  for  the  maximum  voltage  sensitivity  at  light  flux  F,  where  the  load 
resistance  is  equal  to  the  light  resistance  of  the  cell  n,  under  the  condition  of  operation. 
If  the  load  is  a  relay  to  be  operated  directly  in  series  with  the  cell  the  ampere-turn  sensi- 
tivity must  be  used  instead  of  the  above.  This  is  equal  to  the  voltage  sensitivity  given 
above  multiplied  by  1/i,  where  t  is  the  resistance  per  turn  of  the  relay. 

If  a  photoresistance  cell  is  operated  between  dark  and  a  given  light  intensity  as  the 
limits,  the  maximum  voltage  change  across  the  load  resistance,  r«,  is  secured  when 


the  voltage  change  being 


Ttte  same  condition  for  ra  applies  for  the  current-turn  sensitivity,  and  its  change  is  equal 
to  tne  voltage  change  multiplied  by  l/£. 


-l ±_\ 

Ti  -f  ra      r0  -f  ra/ 


BARRIER  PHOTOCELLS 


16-13 


AMPLIFICATION  OF  PHOTO  CONDUCTIVE-CELL  OUTPUT.  The  amplification 
of  the  output  of  these  cells  is  fundamentally  the  same  as  for  the  emissive  cells,  and  the 
same  types  of  amplifiers  can  be  used.  Since  conductive  cells  have  much  lower  impedances, 
it  is  possible  in  many  cases  to  match  the  load  to  the  cell  and  thereby  secure  the  maximum 
efficiency  of  operation.  Since  the  cell  resistance  may  vary  rapidly  with  illumination,  care 
must  be  taken  that  the  match  be  made  at  the  average  light  intensity  at  which  the  cell  is 
to  be  operated.  If  an  intermittent  or  variable  illumination  is  used  such  that  the  cell  must 
respond  to  some  range  of  frequencies  it  may  be  necessary  to  equalize  the  output  in  order 
to  preserve  fidelity  of  reproduction.  Since  there  is  an  increasing  loss  as  the  frequency  of 
response  is  increased,  it  is  necessary  to  compensate  for  it  by  introducing  attenuation  of 
the  lower  frequencies  by  a  filter  network  at  some  convenient  point  in  the  amplifier.  The 
highest  frequency  at  which  one  desires  to  work  will  then  determine  the  effective  loss  of  a 
cell. 

5.  BARRIER  PHOTOCELLS 

STRUCTURE.  The  fact  that,  under  certain  conditions,  photoconductive  selenium 
cells  could  produce  an  emf  on  illumination  without  any  applied  potential  has  been  known 
since  the  late  nineteenth  century.  The  discovery  of  the  effect  in  cuprous  oxide  revived 
interest  in  it  and  led  to  the  development  of  cells  of  commercial  importance.  They  are  of 
the  sandwich  type  of  construction  referred  to  above  and  illustrated  in  Fig.  2(e).  A  photo- 
sensitive material  such  as  selenium  or  cuprous  oxide  C  is  formed  on  a  suitable  metallic 
base  E  and  covered  with  a  translucent  conductor  such  as  a  wire  mesh  A  or  a  thin  metallic 
film.  If  the  photosensitive  layer  is  itself  partially  transparent,  as,  for  example,  cuprous 
oxide,  a  photo  emf  may  appear  at  junction  D,  where  it  is  called  a  "back-wall  effect,"  or 
at  B,  where  it  is  called  a  "front-wall  effect." 
The  location  and  degree  of  sensitiveness  are 
dependent  on  the  method  of  preparation, 
the  heat  treatment,  rate  of  cooling,  gas  con- 
tent, and  treatment  of  the  boundary  sur- 
face. If  an  attempt  is  made  to  pass  current 
across  such  a  photosensitive  boundary  by 
inserting  a  battery  of  a  few  volts  in  the 
meter  circuit,  it  is  customarily  found  that 
the  current  can  flow  much  more  easily  in 
one  direction  than  in  the  other,  and  this  di- 
rectional resistance  behavior  or  rectification 
is  illustrated  in  Fig.  13  for  a  typical  com- 
mercial cell  in  the  dark,  the  unit  here  being 
of  the  front-wall  type.  As  a  photocell  with- 
out the  external  battery,  the  effect  of  illumi- 
nation is  to  make  the  top  become  negative 
and  the  base  positive;  that  is,  the  electrons 
released  by  the  light  flow  in  the  high-resist- 
ance direction.  If  the  active  layer  is  at  the 
back  wall  D,  Fig.  2(e),  the  behavior  is  pri- 
marily the  same  except  that  the  polarities 
are  reversed  with  respect  to  the  top  and  bot- 
tom. In  this  case  the  light  must  penetrate 
much  more  material,  which  reduces  the 
optical  efficiency  and  alters  the  spectral  re- 
sponse curve. 


10000 


9000 


7000 


54000 


3000 


1OOO 


I 


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Top-E 


FIG.    13. 


Dark  Resistance-voltage  Relation  for 
Selenium  Barrier  Cell 


When  a  barrier  cell  is  illuminated,  its  dark  resistance  at  zero  applied  voltage  is  decreased 
by  the  photoconductive  effect  in  the  layer,  and  at  high  intensities  it  may  be  much  reduced 
for  a  cell  of  the  characteristics  shown  in  Fig.  13. 

ILLUMINATION  RESPONSE.  The  current-illumination  response  of  a  typical  cell 
is  shown  in  Fig.  14  with  various  series  resistances.  With  very  low  resistance  or  short- 
circuited  current  the  relation  is  linear  or  very  nearly  so,  gradually  becoming  more  curved 
as  the  resistance  is  increased  until  the  open-circuit  voltage  relation  is  reached.  Figure  15 
shows  the  two  extremes  for  comparison.  Care  must  therefore  be  taken  not  to  use  too 
much  series  resistance  if  a  linear  response  is  desired. 

WAVELENGTH  RESPONSE.  In  Fig.  16,  A  shows  the  spectral  response  for  a  typical 
selenium  cell;  B  and  C  are  for  front-  and  back-wall  cuprous  oxide  cells,  respectively.  In 
general,  cells  of  this  type  have  most  of  their  sensitivity  in  the  visible  region.  Back-wall 
cells  of  cuprous  oxide,  however,  are  deficient  in  this  region  owing  to  the  absorption  of  the 


15-14 


ELECTRO-OPTICAL  DEVICES 


red  oxide  so  that  their  response  is  confined  largely  to  the  visible  red  and  some  distance 
beyond  into  the  infrared.  Cells  equipped  with  optical  filters  to  make  their  response  closely 
that  of  the  eye  are  now  supplied  by  manufacturers. 


40       SO       120     160     200     240     280 
Illumination  in  Foot-candles 

FIG.     14.     Current-illumination     Relation     of 
Barrier  Cell  with  Different  Series  Resistances 


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280 
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Illumination  -  Foot-candles 

Fia    15.     Short-circuit  Current   and   Open- 
circuit     EMF-illumination     Relations      for 
Barrier  Cell 


SENSITIVITY.  Barrier  cells  are  rated  according  to  their  microampere-per-lumen 
output,  and  the  same  qualifications  apply  to  them  as  were  stated  for  emissive  cells  con- 
cerning the  color  of  the  light  source  and  resistance  in  series.  On  account  of  the  warping 
of  the  linearity  of  the  current  curve  by  comparatively  small  resistances,  care  must  be  taken 
in  its  measurement  to  use  a  sufficiently  low-resistance  meter.  It  is  also  frequently  useful 
in  certain  applications  to  have  a  statement  of  the  open-circuit  voltage  in  millivolts  per 
lumen.  From  Fig.  15  it  is  clear  that  this  ratio  is  high  at  low  illuminations  and  rapidly 
diminishes  as  the  illumination  is  raised,  and  for  this  reason  it  is  necessary  to  state  the 
illumination  at  which  the  measurement  is  made. 

Calculations  of  the  maximum  power  and  voltage  sensitivities  are  complicated  by  the 
fact  that  the  internal  resistance  decreases  with  illumination,  especially  with  the  selenium 

type.  For  this  reason,  it  is  necessary 
to  know  the  characteristics  of  the  in- 
dividual cell,  and  it  is  advisable  to 
consult  the  manufacturer  regarding 
the  particular  use  to  which  it  is  to  be 
put  or  to  determine  the  proper  load 
experimentally.  For  the  cuprous  ox- 
ide type  the  internal  resistance  is  com- 
monly assumed  constant,  and  in  this 


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case  the  maximum  voltage  sensitivity, 
dEg/dF,  is  equal  to  Sr,  5  being  the 
sensitivity  constant  and  r  the  internal 
resistance,  the  load,  rz,,  being  rela- 
tively very  large.  In  this  case  the 
maximum  power  sensitivity  is  ^VF/2, 
where  rz,  =  r.  These  values  of  TL  will 
ordinarily  be  sufficient  to  cause  depar- 
ture from  linearity  of  current  response 
so  that  a  compromise  must  be  made  by  sufficient  reduction  in  the  load  resistance.  Barrier 
cells  are  liable  to  deteriorate,  and  direct-reading  instruments  utilizing  them  such  as 
foot-candle  and  photographic  exposure  meters  should  be  checked  occasionally  and  the 
eel  replaced  if  necessary.  Poor  contact  with  the  romping  rings  may  also  develop.  This 


30OO  3500  40OO  4500   5000    55OO  60OO   6500   7000 
Wave-length 

FIG.  16.     Spectral  Response  of  Barrier  Cells.     A,  sele- 
nium.    Br  cuprous  oxide,  front  wall.     C,  cuprous  oxide, 
back  wall 


CHOICE  OF  CELLS  FOR  VARIOUS  PURPOSES         15-15 

type  of  cell  is  subject  to  a  fatigue  which  causes  a  decrease  in  the  response  when  first 
illuminated.  Recovery  takes  place  in  the  dark,  but  the  reproducibility  of  readings  is 
somewhat  dependent  on  the  duration  and  intensity  of  illumination.  This  type  has  no 
dark  current. 

FREQUENCY  RESPONSE.  Exact  information  on  the  various  commercial  cells  is 
not  available.  Barrier  cells  have  a  relatively  large  internal  capacitance  which  diminishes 
their  output  with  increasing  frequency.  The  "Photronic  cell"  is  stated  to  give  a  satis- 
factory response  to  light  interrupted  at  60  cycles  per  second,  and  if  this  output  is  assumed 
100  per  cent  then  at  120  cycles  it  will  be  about  5S  per  cent,  at  240  cycles  30  per  cent, 
and  at  1000  cycles  6.4  per  cent.  It  is  also  stated  that,  if  an  equalized  response  is  produced 
up  to  5000  cycles,  the  power  level  is  reduced  approximately  35  db.  The  response  of  the 
cell  to  single  interruption  is  more  rapid  than  that  of  a  relay,  so  that  for  this  use  they  may 
be  considered  instantaneous. 

AMPLIFYING  CIRCUITS.  If  constant  illumination  is  to  be  used,  a  d-c  amplifier  is 
demanded,  which  may  be  troublesome  to  build  and  operate.  In  general  it  is  therefore 
recommended  that  the  light  be  interrupted  at  a  low  frequency  of,  for  example,  about  60 
cycles,  and  that  a  good  a-f  amplifier  capable  of  transmitting  this  frequency  efficiently 
be  used. 

The  remarks  previously  made  concerning  the  amplification  of  emissive  and  conductive 
cells  also  apply  to  this  type.  Barrier  cells  have  lower  impedance  than  either  of  these,  and 
are  well  within  the  range  of  practical  transformers,  so  that  they  can  be  directly  coupled  and 
the  output  used  in  an  a-c  amplifier.  Equalization  of  frequency  response  may  be  necessary 
as  with  the  conductive  cells. 


6.  PHOTOVOLTAIC  CELLS 

In  the  middle  of  the  nineteenth  century  it  was  discovered  that  if  two  similar  electrodes 
of  certain  materials,  such  as  platinum  or  silver  coated  with  silver  halide,  were  immersed 
in  dilute  electrolytes  and  one  electrode  was  illuminated,  a  voltage  appeared  between  them. 
These  are  referred  to  as  photovoltaic  cells,  although  some  writers  broaden  this  name  to 
include  also  the  barrier  cells  previously  described  and  refer  to  them  as  wet  and  dry  cells 
respectively.  Various  combinations  of  electrodes,  coatings,  and  intervening  liquids  have 
been  employed,  but,  apart  from  experimental  studies  of  the  effect,  the  usual  materials  are 
oxides,  sulfides,  or  halides  in  an  acid  or  inorganic  salt  electrolyte.  Cuprous  oxide  gives  a 
large  effect  and  has  frequently  been  used  in  attempts  to  commercialize  this  type,  the  non- 
sensitive  electrode  being  some  durable  material  such  as  lead  with  an  electrolyte  of  lead 
nitrate  in  water;  see  Fig.  2(/). 

CHARACTERISTICS.  The  behavior  of  the  open-circuit  voltage  and  the  short-circuit 
current  responses  with  illumination  for  a  cell  of  the  Cu,  Cu^O,  PbCNOaK  Pb  construction 
are  similar  in  shape  to  the  corresponding  characteristics  for  barrier  cells  (see  Fig.  15) ,  the 
former  showing  a  tendency  to  voltage  saturation  and  the  latter  being  linear.  The  effect 
of  increasing  external  resistance  is  also  qualitatively  similar;  see  Fig.  14. 

The  spectral  response  of  the  cuprous  oxide  cell  is  high  in  the  visible  spectrum,  being  a 
maximum  in  the  blue  or  blue-green. 

These  cells  are  subject  to  polarizing  effects  which  make  them  more  unstable  than  the 
other  types,  and  there  is  a  gradual  deterioration  of  the  sensitive  layer  which  greatly 
shortens  then*  useful  life.  It  is  claimed  that  this  deterioration  can  be  inhibited  to  some 
extent  by  using  a  depolarizer  such  as  hydrogen  peroxide  in  the  electrolyte  to  oxidize  the 
free  hydrogen  which  reduces  the  cuprous  oxide. 

The  response  to  illumination  and  the  recovery  afterward  are  not  as  rapid  as  the  cuprous 
oxide  barrier  effect. 

7.  CHOICE  OF  CELLS  FOR  VARIOUS  PURPOSES 

Although  the  applications  of  photocells  are  frequently  classified  according  to  marginal 
or  linear  operation,  this  does  not  have  so  much  to  do  with  the  choice  of  the  cell  as  the 
method  of  its  use  and  of  the  amplification  of  its  output.  Ordinarily  the  selection  of  a  cell 
is  dominated  by  considerations  of  high  sensitivity  to  tungsten  or  daylight,  the  requirements 
of  precision  photometry,  a  particular  spectral  response,  or  convenience,  all  of  which 
involve  the  relative  evaluation  for  the  purpose  at  hand  of  such  factors  as  magnitude  of 
output,  frequency  loss,  fidelity  of  color  and  intensity  response,  permanence,  stability, 
leakage,  and  absence  of  external  battery. 


15-16 


ELECTRO-OPTICAL  DEVICES 


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CHOICE   OF  CELLS  FOR  VARIOUS  PURPOSES         15-17 


Rauland 
Westinghouse 
Westinghouso 
G.E.  Co. 
G,E.  Co. 

§§§! 

Weston 
Weston 
Selenium  Corp. 
G.E.  Co. 
G.E.  Co. 

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15-18  ELECTRO-OPTICAL  DEVICES 

The  cesium  oxide  cell  is  particularly  efficient  for  use  with,  tungsten  light  because  its 
maximum  sensitivity  is  near  the  optimum  energy  emission  of  the  high-brilliancy  gas 
lamps.  This  renders  it  suitable  for  sound  pictures,  telephotography,  television,  and  numer- 
ous marginal  applications  such  as  counting  and  control  operations.  The  color  response 
is  also  sufficiently  extended  over  the  spectrum  to  permit  its  use  in  many  sorting  operations 
and  in  colorimeters.  In  some  technical  applications  it  is  necessary  or,  at  least,  highly 
desirable  that  the  cells  be  used  with  infrared  light  to  avoid  detection  of  the  beam  by  the 
wary  or  curious,  and  the  cell  has  sufficient  infrared  response  to  permit  the  tungsten  light 
to  be  concealed  by  filters  which  transmit  only  this  region  with  very  little  visibility  of  the 
interrupted  beam.  The  cesium-antimony  cathode  is  rinding  increasing  application.  Its 
high  efficiency  to  tungsten  light  makes  it  competitive  with  cesium  oxide  and  considerably 
superior  with  bluer  modern  illuminants.  For  colorimetric  applications,  a  surface  such  as 
the  S3  is  more  suitable  because  of  its  broad  spectral  response  through  the  visible  region 
which  more  nearly  simulates  the  eye. 

The  requirements  of  precision  photometry  vary  greatly  according  to  the  intensity  and 
color  of  the  light  to  be  measured  and  according  to  whether  the  measurements  are  relative 
or  in  visual  units.  For  example,  in  stellar  photometry  where  very  little  light  is  available 
and  high  amplification  is  necessary,  leakage  in  and  around  the  cell  must  be  reduced  to  the 
minimum  and  high  sensitivity  and  color  response  may  not  be  as  important.  On  the  other 
hand,  in  the  ordinary  routine  photometry  of  tungsten  light,  leakage  may  be  relatively 
unimportant.  In  precision  photometry  it  is  essential  that  cells  be  used  in  a  manner  which 
gives  an  assured  calibration  for  each  reading  and  that  their  constancy  be  not  assumed 
without  adequate  proof.  Cells  to  be  used  for  measurements  in  visibility  units  require 
either  a  filter  to  modify  their  color  response  or  a  calibration  by  a  source  whose  spectral 
emission  is  the  same  as  that  to  be  measured.  If  sources  such  as  tungsten  lamps  through  a 
moderate  range  of  color  temperature  are  to  be  measured,  requirements  of  commercial 
photometry  may  be  met  by  calibrating  with  a  similar  source  within  the  range,  combined 
if  necessary  with  a  visual  filter  that  approximately  matches  the  eye. 

The  increasing  use  of  ultraviolet  for  therapeutic  and  photochemical  purposes  has  created 
a  demand  for  cells  to  measure  intensities  in  this  region,  the  response  for  therapeutic  pur- 
poses being  of  such  shape  as  to  evaluate  the  radiation  directly  in  erythema  units  or  in 
time  of  exposure.  This  region  is  from  2SOO  to  3200  angstroms  approximately,  the  shorter 
wavelengths  of  questionable  value  being  excluded  by  filtering  the  source.  A  number  of 
materials  are  intrinsically  sensitive  to  this  region,  but  it  is  undesirable  to  use  those  of 
considerable  visible  sensitivity  because  of  the  difficulty  of  suppressing  sufficiently  the 
larger  amount  of  energy  in  the  longer  wavelengths  even  in  the  mercury  arc,  which  would 
mask  the  ultraviolet  response.  Consequently  it  is  desirable  to  use  only  those  materials 
which  are  not  naturally  sensitive  much  beyond  3200  angstroms.  Cadmium,  uranium,  and 
lithium  have  been  proposed,  and  the  first  two  have  been  used  practically.  In  order  to 
limit  the  response  to  the  proper  wavelength  on  the  short-wavelength  side,  these  metals 
can  be  mounted  in  bulbs  of  Corex  D  glass  instead  of  quartz.  If,  for  other  purposes,  it  is 
desired  to  broaden  the  region  of  response  up  toward  or  into  the  visible  blue,  thorium  and 
cerium  have  been  suggested. 

The  glass  bulbs  of  ordinary  commercial  cells  are  more  or  less  opaque  to  radiation  beyond 
about  3200  angstroms,  so  that,  hi  any  event,  a  special  glass  or  quartz  is  necessary.  The 
cesium  oxide  cathode,  however,  is  very  sensitive  to  the  longer  ultraviolet  up  to  2000  ang- 
stroms as  far  as  measurements  are  available. 

For  certain  purposes  where  an  external  battery  is  undesirable,  cells  of  the  barrier  type 
are  finding  application,  an  example  being  as  a  photographic  exposure  meter.  It  is  neces- 
sary, of  course,  to  interpret  the  readings  in  terms  of  exposures  for  the  various  color  sensi- 
tivities of  emulsions  by  charts  or  suitable  scales  on  the  meter. 

Table  3  lists  the  better-known  commercial  cells. 

BIBLIOGRAPHY 

Gudden,  LichfeleMri&che  ErscJieinungen,  Julius  Springer,  Berlin,  192ST 

CiTnpbell  and  Ritchie,  Photoelectric  Cells,  Sir  Isaac  Pitman  and  Sons,  1929. 

Sinaon  and  Stihrman,  Lichtelektrische  Zellen,  Julius  Springer,  Berlin,  1932. 

Fleischer  and  Teiehman,  Die  lichtelektrische  Zelle,  Theodor  Steinkopff,  Dresden  and  Leipzig,  1932. 

Hughes  and  DuBridge,  Photoelectric  Phenomena,  McGraw-Hill  Book  Co.   1932. 

Barnard,  The  Selenium  Cell,  Richard  R.  Smith,  New  York,  1930. 


Deutsch- 


Zworykin  and  Wilson,  Photocells  and  Their  Applications,  John  Wiley  <fe  Sons,  1932 
Doty,  Selenium,  A  List  of  References,  1817-19B5,  New  York  Public  Library,  1927. 
Walker  and  Lance,  Photoelectric  Cell  Applications,  Pitman,  London,  1933. 


THE  IMAGE  DISSECTOR  15-19 

Handbuch  der  Physik,  Vol.  XIX,  pp.  829  ff.     (Thermal  cells.) 

Koller,  Physics  of  Electron  Tubes,  McGraw-Hill  Book  Co.,  New  York  and  London,  1937. 
Henney,  Electron  Tubes  in  Industry,  McGraw-Hill  Book  Co.,  New  York  and  London,  1937. 
Reich,  Theory  and  Applications  of  Electron  Tubes.  McGraw-Hill  Book  Co.,  New  York  and  London,  1944. 
'  ~     '        ""         -         -  -  -    ~     -    -  -  354  (June  1946)> 

35S  (June  1946). 


Brattain  and  Becker,  Thermistor  Bolometers,  J.  Optical  Soc.  Am.,  Vol.  36,  6,  p.  354  (June  1946). 
Cashman,  R.  J.,  New  Photoconductive  Cells,  /.  Optical  Soc.  Am.,  Vol.  36,  6,  p.  ?~~   "         


TELEVISION  PICK-UP  TUBES 

By  V.  K.  Zworykin  and  E.  G.  Ramberg 

8.  REQUIREMENTS 

The  purpose  of  the  television  pick-up  tube  is  to  convert  an  optical  image  of  the  scene 
to  be  transmitted  into  an  electrical  signal  descriptive  of  the  light  distribution  in  the  image. 
In.  all  the  pick-up  tubes  here  considered  the  signal  is  obtained  by  scanning  in  sequence  a 
rectangular  image  area  along  a  fixed  number  (e.g.,  525)  of  adjoining  horizontal  scanning 
lines;  with  an  ideal  transmission  system  and  viewing  device  the  instantaneous  signal  output 
of  the  pick-up  tube  determines  the  brightness  of  a  particular  picture  element  (i.e.,  a  square 
whose  length  and  height  are  equal  to  the  separation  of  two  scanning  lines)  in  the  reproduced 
image,  scanned  in  synchronism  with  the  transmitted  image. 

A  satisfactory  pick-up  tube  must  be  capable  of  furnishing  a  signal  that  can  be  converted 
into  an  image  with  adequate  detail,  free  from  objectionable  random  fluctuations  in  bright- 
ness (noise) ,  and  faithful  in  geometry  and  tonal  values  over  its  entire  area,  at  a  reasonable 
illumination  of  the  transmitted  scene.  Similar  requirements  regarding  resolution,  signal- 
to-noise  ratio,  uniformity,  and  sensitivity  must  also  be  fulfilled  by  the  35-mm  negative 
film  employed  in  commercial  motion-picture  production,  whose  properties  may  reasonably 
be  taken  as  a  standard.  This  is  all  the  more  appropriate  since  the  comparison  of  television 
with  motion  pictures  appears  inescapable. 

SENSITIVITY.  A  suitable  figure  of  merit  for  the  sensitivity  is  given  by  the  ratio 
/Y  (BA) ,  where  /is  the  /-number  of  the  lens  employed,  B  the  brightness  of  the  scene  required 
to  yield  a  good  picture  (in  lumens  per  square  meter) ,  and  A  the  area  of  the  picture  on  the 
film  (in  square  meters) ;  through  the  factor  f/A  the  figure  of  merit  is  proportional  to  the 
square  of  the  depth  of  focus.  Employing  figures  derived  from  motion-picture  studio 
practice,  /  =  2,  B  =  5500  (lumens/m2),  and  A  =  0.00032  m2  (0.5  in.*),  J*/{BA)  =  2.3. 

RESOLUTION  AND  SIGNAL-TO-NOISE  RATIO.  Thirty-five-millimeter  film  is 
generally  capable  of  resolving  1000  to  1500  lines  per  picture  height;  on  the  other  hand, 
at  this  level  the  photographic  grain  or  noise  interferes  seriously  with  the  picture  detail 
(i.e.,  the  signal).  For  a  ratio  of  the  signal  to  the  root-mean-square  noise  amplitude  of 
30-40,  required  to  render  the  grain  unobjectionable,  the  resolution  must  be  reduced  to 
about  500  lines.  It  should  be  noted  that  the  root-mean-square  noise  amplitude  employed 
throughout  in  the  present  discussion  is  only  about  one-sixth  as  great  as  the  peak-to-peak 
noise  amplitude,  which  may  be  observed  directly  on  an  oscilloscope  screen.  It  is  found 
experimentally  that  the  signal-to-noise  ratio  for  film  remains  approximately  constant 
throughout  the  useful  exposure  range.  It  differs  in  this  from  the  more  sensitive  television 
pick-up  tubes,  for  which  the  noise  is  constant  and  the  signal-to-noise  ratio,  hence,  is  lower 
in  the  low  lights  than  in  the  high  lights. 

UNIFORMITY.  The  W™  image  is  geometrically  faithful  and  uniform  in  response  over 
the  entire  image  area. 

It  will  be  seen  that  certain  pick-up  tubes  exhibit  higher  sensitivity  and  signal-to-noise 
ratio  for  equal  resolution  than  film.  To  this  extent  they  enable  television  cameras  to 
function  more  favorably  than  studio  and  news  motion-picture  cameras. 

9.  THE  IMAGE  DISSECTOR 

The  Farnsworth  image  dissector  is  shown,  in  schematic  cross-section,  in  Fig.  1.  At  one 
end  of  the  tube  there  is  a  photocathode  on  which  a  lens  projects  an  optical  image  of  the 
scene;  at  the  other,  a  positive  electrode  with  a  tiny  aperture,  equal  to  a  picture  element 
in  size.  The  magnetic  field  of  a  solenoid  focuses  the  photoelectrons  so  as  to  form,  in  the 
plane  of  the  aperture,  a  charge  image  of  the  picture  on  the  photocathode.  This  charge 
image  is  swept  across  the  aperture  by  the  magnetic  deflecting  fields  so  that,  at  any  instant, 
photoelectrons  from  just  one  picture  element  on  the  photocathode  pass  through  the 
aperture.  These  accelerated  photoelectrons  fall  on  the  first  stage  of  an  11-stage  multiplier 
(see  article  4)  built  into  the  tube  and  eject  a  larger  number  of  secondary  electrons  which 


15-20 


ELECTRO-OPTICAL  DEVICES 


are  drawn  through,  an  accelerating  screen  to  the  second  target  electrode,  leading  to  a 
further  secondary-emission  multiplication  of  the  current.  The  output  of  the  multiplier, 
finally,  may  be  coupled  by  a  resistance  to  the  input  of  a  standard  video  amplifier.  _ 


.Magnetic  focusing  coll 


"ZTEfeotron  multiplier 
..Nickef  wall  coating 


Photo  cathode 


-  Output  lead 
FIG.  1.     The  Image  Dissector 

The  current  passing  through  the  aperture  is  simply  the  photocurrent  emitted  by  a 
picture  element  on  the  photocathode.  Thus,  if  p  is  the  photosensitivity  of  the  cathode 
in  amperes  per  lumen,  L  its  illumination  in  lux  Gumen/m2),  A  the  effective  area  of  the 
photocathode  in  square  meters,  and  N  the  number  of  picture  elements,  the  signal  current 
becomes 

^Denoting  the  transmitted  band  width  by  F,  the  shot-noise  amplitude  for  this  current  is 
Assuming  F  =  5  -  106  sec"1  and  a  525-line  picture  (2V  =  (525)2  -  4/3), 


^2x2  —  2.1  • 
Hence  the  signal-to-noise  ratio  is 

c        jf  I  *  *yz         i  o 
w  —  tg'  Trn         —    I  .o 

If  it  is  assumed  that  p  =  20  *  10  ~*  amp/lumen,  A  —  0.01  m2  (15  in.2) 


(3) 
(4) 

(5) 

A  signal-to-noise  ratio  of  100  would  thus  demand  a  cathode  illumination  L  =  29,000  lux. 

The  multiplication  provided  by  the  multiplier  should  be  such  that  the  multiplied  shot 

noise  exceeds  the  amplifier  input  tube  noise  current,  which  may  be  estimated  at  2  •  10  ~9 

amp.    Since  

i^  -  1.6  -  10-»S  (6) 

the  multiplier  gain  will  suffice  for  all  recognizable  picture  detail  (S  >  1)  if  it  is  equal  to  a 
few  thousand.  The  actual  gain  is  made  larger  than,  this,  reducing  the  required  amplifier  gain. 
SENSITIVITY.  From  the  above  figures  it  follows  that,  in  order  to  transmit  a  picture 
with  a  signal-to-noise  ratio  of  100,  an  image  dissector  provided  with  an  //4.5  lens  would 
require  (for  an  effective  cathode  area  of  0.01  m2)  a  high-light  brightness  of  the  scene  equal 
to  29,000  (2-4.S)2  =  2.3 -106  lumens/m2.  The  figure  of  merit  of  the  dissector,  calculated 

in  the  same  manner  as  for 


D'fssector 


film,  hence  becomes 
4.52  1 

0.01 -2.3- 105  ~  1100 


(7) 


FIG.  2.     Motion  Picture  Transmission  with  Image  Dissector 


Thus  the  sensitivity  of  the 
dissector  is  less  than  that  of 
film  by  a  factor  of  two  or 
three  thousand.  This  is  a 
drawback  for  direct  pick-up 
but  is  of  secondary  import- 
ance for  motion-picture 


2). 


For  the  latter  purpose,  continuously  moving  film  is  generally  employed 
Ttie  film  motion  provides  the  vertical  deflection,  so  that  only  the  horizontal 
coil  need  be  actuated.  Special  types  of  image  dissectors,  designed  spe<jifically 


THE  ICONOSCOPE 


15-21 


for  the  transmission  of  black  and  white  and  of  color  film,  are  available.  Since  the  standard 
projection  speed  for  motion  pictures  is  24  frames  per  second  and  the  television  field 
frequency  is  60  per  second,  the  arrangement  shown  in  Fig.  2  is  generally  modified  by 
the  insertion  of  an  optical  compensation  system  between  the  continuously  moving  film 
and  the  dissector  tube.  This  causes  the  motion-picture  frames  to  be  scanned  two  and 
three  times  in  alternation. 

RESOLUTION.  Under  normal  circumstances  the  aperture  size  determines  the  resolu- 
tion of  the  image  dissector.  Change  of  focus  with  deflection  is  readily  compensated  by 
applying  the  proper  correcting  signals  to  the  focusing  current  or  the  accelerating  voltage 
in  synchronism  with  the  deflection.  Fundamentally,  the  resolution  of  this  type  of  tube 
is  limited  by  the  unsharpness  of  focus  arising  from  the  initial  velocities  of  the  photo- 
electrons.  An  increase  in  the  number  of  picture  elements  demands,  hence,  either  the  em- 
ployment of  a  larger  photocathode,  leaving  the  length  of  the  tube  unaltered,  or  a  higher 
operating  voltage  and  stronger  focusing  field. 

SIGNAL-TO-NOISE  RATIO.  The  signal-to-noise  ratio  of  the  image  dissector  is  pro- 
portional to  the  square  root  of  the  scene  brightness.  Hence,  for  equal  signal-to-noise  ratio 
in  the  high  lights,  the  noise  will  be  more  prominent  in  the  low  lights  than  for  film,  though 
less  prominent  than  for  most  of  the  remaining  pick-up  tubes  to  be  considered,  for  which  the 
noise  is  independent  of  the  light  level.  It  is  therefore  necessary  to  demand  a  higher  signal- 
to-noise  ratio  in  the  high  lights  than  for  film  (e.g.,  100  in  place  of  30-40).  The  fact  that 
the  noise  becomes  more  noticeable  in  the  low  lights  is  accentuated  by  the  circumstance 
that  the  signal  output  of  the  dissector  is  strictly  proportional  to  the  element  brightness, 
i.e.,  that  the  tonal  scale  is  not  compressed  by  the  pick-up  device. 

UNIFORMITY.  The  image  dissector  has  excellent  uniformity  properties,  both  with 
regard  to  constancy  of  response  over  the  entire  picture  and  to  the  absence  of  geometric 
distortions. 

10.  THE  ICONOSCOPE 

The  iconoscope,  the  orthicon,  and  the  image  orthicon  may  be  classed  together  as  storage 
pick-up  tubes.  In  all  of  them  the  charge  released  photoelectrically  from  a  picture  element 
by  the  incident  light  is  stored  in  the  period  intervening  between  two  successive  scannings 
of  the  element.  This  leads  to  a  very  great  gain  in  sensitivity  in  comparison  with  non- 
storage  pick-up  systems  such  as  the  image  dissector. 

Figure  3  shows  the  construction  of  a  standard  iconoscope  (type  1850-A)  with  magnetic 
deflection.  It  is  seen  to  consist  of  an  electron  gun  whose  beam  is  deflected  across  the 
surface  of  a  photosensitive 
"mosaic"  by  two  pairs  of  Mosaic 
external  deflecting  coils,  and 
the  mosaic,  all  enclosed  in 
a  dipper-shaped  envelope. 
An  image  of  the  scene  to 
be  transmitted  is  projected 
by  a  lens  through  an  opti- 
cally clear  face  of  the  enve- 
lope onto  the  mosaic  plate 
whose  dimensions  are  4  3/4 
X  3  9/i6  in.2 

A  typical  gun  structure 
consists  of  an  indirectly 
heated  cathode  enclosed  in 
a  grid  cylinder  with  a  0.040- 
in.  aperture,  a  closely  spaced 
cylindrical  accelerating  elec- 
trode at  full  anode  voltage 
with  a  denning  aperture  0.002  in.  in  diameter,  a  focusing  electrode  or  first  anode,  and 
the  final  anode  in  the  form  of  a  platinum  coating  on  the  inner  wall  of  the  gun  tube. 
The  defining  aperture,  which  coincides  approximately  with  the  cross-over,  is  imaged  by  the 
equipotential  electron  lens  formed  by  the  two  cylinders  at  full  anode  potential  and  the 
intermediate  focusing  electrode  on  the  mosaic,  forming  a  spot  0.005  in.  or  less  in  diameter; 
this  design  minimizes  the  current  striking  the  focusing  electrode  or  first  anode  and  hence 
prevents  disturbing  secondary  emission  from  the  gun.  In  practice  the  anode  is  maintained 
at  approximately  1000  volts,  the  focusing  electrode  at  300  volts,  and  the  grid  bias  may  be 
varied  from  —  30  to  —  50  volts.  The  optimum  Iconoscope  beam  current  ranges  from  0.05 
to  0.2  microampere,  increasing  with  the  illumination  of  the  mosaic. 


v       ^-~ 

Magnetic  deflection 

Focusing  electrode 
~  (1st  anode) 


Grid 


PreampiJEei 


The  Iconoscope 


15-22 


ELECTRO-OPTICAL  DEVICES 


100 


The  mosaic  is  a  thin  sheet  of  mica,  covered,  on  the  side  facing  the  lens  and  the  electron 
beam,  with  an  array  of  minute  silver  globules,  small  compared  with  a  picture  element. 
These  have  been  rendered  photosensitive  by  a  process  involving  oxidation,  cesiation,  and 
the  subsequent  evaporation  of  silver.  On  the  other  side  the  mica  sheet  is  coated  with  a 

continuous  metal  film,  the  signal  plate,  which 
is  electrically  connected  to  the  coupling  re- 
sistor and  the  grid  of  the  first  stage  of  ampli- 
fication. The  capacitance  between  the  signal 
plate  and  the  photosensitive  mosaic  is  of  the 
order  of  1  pcf/m2;  the  total  capacitance  be- 
tween signal  plate  and  anode  coating,  10  /t/zf. 
The  photosensitivity  of  the  mosaic  is  4-10  jua/ 
lumen;  its  spectral  response  is  shown  in  Fig. 
4.  For  a  mosaic  illumination  of  10  to  50  lux 
a  coupling  resistance  of  0.1  megohm  is  recom- 
mended. At  very  low  light  levels  (and  a 
beam  current  of  the  order  of  0.05  ju&)  it  is 
proper  to  increase  this  to  1  megohm. 

Figure  5  shows  a  smaller  Iconoscope  (type 
5527) ,  with  electrostatic  deflection  and  trans- 
parent signal  plate,  which  is  designed  pri- 
marily for  industrial  and  amateur  use.  It 
has  a  1.4-in.  mosaic  and  operates  with  a 
beam  voltage  of  800  volts  and  a  first-anode 
voltage  between  125  and  250  volts.  The 
cut-off  voltage  for  the  control  grid  is  about 
—  75  volts,  and  the  horizontal  and  vertical 
deflection  voltages  are  in  the  neighborhood  of 
100  volts. 


3000 


4OOO  5000  6000 

Wavelength,  angstroms 


7000 


FIG.  4.     Spectral  Response  of  the  Type  1850-A 
Iconoscope 


OPERATION,  Superficially,  the  operation  of  the  Iconoscope  may  be  described  as 
follows:  The  mosaic  functions  as  an  array  of  minute  photoelectric  cells  with  a  common 
anode,  whose  cathodes  are  capacitatively  coupled  to  the  signal  plate.  The  elementary 
condensers  so  formed  charge  up,  as  the  mosaic  is  exposed  to  light,  by  an  amount  propor- 
tional to  the  light  intensity.  Whenever  the  beam,  acting  as  a  commutator,  sweeps  across 
them,  the  cathode  elements  are  returned  to  their  equilibrium  potential  by  collecting  the 
requisite  number  of  electrons  from  the  beam,  and  an  equal  electron  current  passes  through 


plates 


FIG.  5.     Small  Iconoscope  with  Electrostatic  Deflection  (Type  5527) 

the  signal  lead.     For  an  illuminated  element  on  a  generally  dark  mosaic  (for  which  the 
total  photocurrent  would  be  very  small),  the  signal  current  would  be  given  by 

AT 


'J  TT  =  PLA 


(S) 


employing  the  following  notation: 

p     photosensitivity  of  mosaic. 

L     luminous  flux/unit  area  of  mosaic. 

.4     area  of  mosaic. 


N    number  of  picture  elements. 
T    frame  time. 

*e     time  required  to  sweep  over  one  picture 
element. 

This  would  represent  a  gain  in  sensitivity  relative  to  non-storage  devices  (without  second- 
ary-emission multiplication  of  the  signal)  which  is  equal  to  the  number  of  picture  elements. 
The  above  representation  is  greatly  oversimplified.  Thus,  it  assumes  that  a  mosaic 
soanped  in  darkness  has  a  uniform  potential.  The  actual,  measured,  potential  distribution, 
at  the  moment  when  the  scanning  beam  is  approximately  a  third  from  the  top  of  the 


THE  ICONOSCOPE 


15-23 


-t- 


Top  of  mosaic 


mosaic,  is  as  represented  in  Fig.  6.  Figure  7  shows  the  variation  of  the  potential  of  a 
particular  illuminated  and  unilluminated  element  in  the  course  of  a  frame  time.  This 
behavior  arises  in  the  following  manner.  Considering  the  unilluminated  mosaic,  an  element 
directly  under  the  beam  emits  secondary  electrons  whose  initial  kinetic  energy  varies  from 
zero  to  a  few  electron  volts.  The  secondary-emission  properties  of  the  mosaic  are  such 
that,  on  the  average,  about  4  secondary  electrons  are  emitted  for  every  primary  electron 
incident  from  the  beam.  These  will  be  able  to  leave  the  element  only  if  the  field  conditions 
in  front  of  it  are  favorable;  as  the  element  becomes,  as  the  result  of  secondary  emission, 
more  positive  with  respect  to  the  anode  and  the  remainder  of  the  mosaic,  a  larger  propor- 
tion of  the  electrons  will  return  to  the  element.  When  the  element  reaches  a  potential  V\ 
of  the  order  of  3  volts  positive  with  respect  to  the 
anode  coating,  only  one  secondary  electron  will  leave 
the  element  for  every  incident  beam  electron  and  no 
further  charging  will  take  place.  The  beam  current 
is  chosen  large  enough  to  bring  the  picture  element 
to  the  equilibrium  potential  Vi  in  every  transit. 

Since  the  mosaic  is  insulated,  on  the  average  only 
one  of  all  the  secondary  electrons  (and  photoelectrons) 
which  leave  the  element  for  every  incident  beam  elec- 
tron arrives  at  the  second  anode;  the  rest  are  redis- 
tributed over  the  remainder  of  the  mosaic.  This  re- 
distribution is  influenced  by  the  potential  distribution 
over  the  mosaic  (and  hence,  for  an  illuminated  mosaic, 
also,  to  some  extent,  by  the  light  distribution  in  the 
image)  and  the  geometry  of  the  tube — in  particular 
the  location  of  clear  glass  surfaces  relative  to  the 
mosaic.  The  redistribution  quickly  reduces  the  po- 
tential of  the  elements  immediately  behind  the  beam 
and  more  gradually  that  of  the  more  remote  elements, 


-KL      +2 


OB 

j! 


+3 
Volts 


Bottom  of  mosaic 


FIG.  6.     Voltage  Variation  over  Icono- 
scope Mosaic  Scanned  in  Darkness 

When  the&elements  have  attained  a  potential  Vz,  of  the  order  of  - 1  1/2  volts,  no  further 
redistributed  electrons  reach  them.  This,  thus,  represents  the  equilibrium  potential  of 
elements  not  under  the  beam. 

Consider,  next,  an  illuminated  element  of  the  mosaic.  Immediately  after  the  beam  has 
rendered  the  element  about  3  volts  positive  with  respect  to  the  second  anode,  no  photo- 
electrons  are  able  to  reach  the  second  anode;  however,  an  appreciable  number  may  find 
their  way  to  the  elements  ahead  of  it  which  have  been  under  the  beam  even  more  recently 
and  hence  are  more  positive.  Thus  the  element  becomes  negative  less  rapidly  than  an 
unilluminated  element.  However,  the  photoemission  is  far  from  saturated,  a  large  propor- 
tion of  the  photocurrent  returning  to  the  element  of  origin.  Although  a  larger  proportion 
of  the  photocurrent  will  reach  the  anode  as  the  element  becomes  more  negative,  the  condi- 
tion of  incomplete  saturation  generally  persists  practically  up  to  the  succeeding  ^passage 
of  the  scanning  beam,  at  which  point  the  illuminated  element  may  have  a  potential  PS,  a 
fraction  of  a  volt  above  F2.  At  low  light  levels  the  average  photocurrent  leaving  a  small 

illuminated  region,  with  the  rest  of 
the  mosaic  ha  darkness,  is  approxi- 
mately 20  per  cent  of  the  saturated 
photoemission. 

As  the  beam  passes  over  the  illu- 
minated element,  it  returns  it  to  the 
positive  equilibrium  potential  "Fi. 
For  a  small  illuminated  area  on  a, 
dark  background,  the  signal  current 
is  equal  to  the  difference  in  the  frac- 
tion of  the  secondary-emission  cur- 


FIG.  7.     Voltage  Variation  of  an  Uliiminated  and  an 
illuminated  Picture  Element  on  the  Iconoscope  Mosaic 


rent  from  the  element  which  reaches  the  anode  for  the  illuminated  and  for  an  unillumi- 
nated region.  This  is  simply  the  current  reaching  the  anode  as  an  uniBuminated  element 
is  raised,  under  the  beam,  from  the  potential  F2  to  the  potential  F3  (see  Fig.  7).  Since 
for  both  these  potentials  the  secondary  emission  is,  in  general,  saturated,  and  the  current 
collected  by  the  anode  must  equal  the  beam  current,  only  a  fourth  of  the  stored  charge 
can  be  utilized  for  the  signal  current  if  the  secondary-emission  ratio  is  4.  Thus  the  total 
operating  efficiency  at  low  light  levels  is  of  the  order  of  5  per  cent  (1/4  of  20  per  cent), 
and  the  signal  current  is  given  by 

i,  =  kpLA        k  ^  0.05  (9) 

It  may  be  noted  that  the  photoelectric  efficiency  can  be  improved  appreciably  by  iHuml- 


15-24 


ELECTRO-OPTICAL  DEVICES 


nating  slightly  the  photosensitive  clear  glass  walls  of  the  tube  (backlighting),  since  this 
raises  their  potential. 

At  high  light  values  the  efficiency  becomes  much  less;  regardless  of  the  degree  of  illumi- 
nation, photoemission  will  drive  an  illuminated  region  only  positive  enough  relative  to 
its  surroundings  to  prevent  the  departure  of  additional  photoelectrons.  Thus  the  Icono- 
scope signal  is  compressed  in  the  high  lights.  The  preferred  collection  of  the  redistributed 
electrons  by  the  more  positive  areas  of  the  mosaic  enhances  this  effect.  At  very  high  light 
levels  the  Iconoscope  signal  is  determined  by  the  photoelectric  charge  stored  during  the 
line  scan  of  the  beam  preceding  the  scanning  of  the  picture  element  considered:  Since  under 
the  beam  an  element  becomes  positive  by  3  volts  relative  to  the  second  anode,  the  photo- 
emission  of  the  neighboring  element  on 
the  Line  ahead  of  the  scanned  line  is 
saturated  even  if  it  is  positive  by  sev- 
eral volts  relative  to  its  other  neigh- 
bors ("line  sensitivity") ;  an  opposing 
voltage  of  1-2  volts  suffices  to  suppress 
the  photoemission. 

The  signal  output  characteristic  of 
the  1850-A  Iconoscope  is  shown  in 
Fig.  8. 

SIGNAL-TO-NOISE  RATIO.  The 
principal  source  of  noise  in  the  conven- 
tional Iconoscope  pick-up  system  is 
the  noise  introduced  by  the  input  tube 
of  the  signal  amplifier,  which  may  be 
represented  as  thermal  noise  from  an 
equivalent  resistor  TT  added  to  the  re- 
sistive component  of  the  coupling  net- 
work between  the  Iconoscope  and  the 
amplifier.  Since,  for  coupling  resist- 
ances of  the  order  of  0.1  to  1  megohm, 
the  coupling  impedance  is  primarily 
capacitative  for  the  high  frequencies 
of  the  video  band  (the  combined  ca- 
pacitance to  ground  of  the  signal  plate 
and  the  grid  of  the  input  tube  may  be  of  the  order  of  15  pid,  corresponding  to  about 
2000  ohrns  at  5  megacycles),  it  is  necessary  to  insert  a  peaking  network  in  the  amplifier 
to  equalize  response  at  low  and  high  frequencies.  This  network  causes  the  noise  spectrum 
to  be  concentrated  in  the  high  frequencies.  For  low-noise  input  tubes  and  a  video  band 
of  5  megacycles,  the  ratio  of  the  signal  to  the  amplitude  of  the  integrated  noise  may  be 
calculated  to  be 

S  =  5-108i«  (10) 

4  being  measured  in  amperes.  It  should  be  noted  that  about  2  or  3  times  as  much  of  this 
peaked  noise  can  be  tolerated  by  the  observer  as  shot  noise  (e.g.,  from  a  multiplier)  dis- 
tributed uniformly  over  the  spectrum. 

SENSITIVITY.  In  practice  a  scene  brightness  from  6000  to  17,000  lumen/ma  is  found 
to  give  satisfactory  pictures  with  an  1850-A  Iconoscope  used  in  conjunction  with  a  lens 
of  //5.6  or  smaller  aperture.  The  target  area  of  the  mosaic  is  approximately  0.011  m2 
(17  in.2).  Employing  the  lowest  values  both  for  the  illiiminati.on  and  the /-number,  the 
figure  of  merit  used  as  a  measure  of  the  sensitivity  becomes 


microamperes 

0  0  C 

M  to  i 

Ol  O  C 

/ 

""" 

/ 

' 

/ 

/ 

Signal  output 
p  p 

to  01  o 

/ 

/ 

' 

5       7      10            2O      30  40  50    70   100           20 
Highlight  illumination  of  mosaic,  lux 

FIG.    8.     SIgnal-versus-Light    Characteristic    of    Tyi 
1850-A  Iconoscope 

5.62 


6000-0.011 


0.48 


(11) 


This  is  about  J/5  that  for  film;  the  formula  for  the  signal-to-noise  ratio  yields  a  value  of 
100,  in  harmony  with  direct  measurements.  This  is  greater  than  for  filrr>  by  a  factor  be- 
tween 2  and  3.  Such  a  factor  is  needed  to  render  the  noise  unobjectionable  in  the  low 
lights,  since  the  noise  does  not  decrease  in  proportion  with  the  signal.  The  fact  that  the 
signal  output  of  the  Iconoscope  has  less  contrast  than  the  original  image,  i.e., 


1  AT. 

7  —    where  k  -  2  —  3 

k  L 


(12) 


a  condition  which  is  generally  compensated  in  the  viewing  tube,  leads  to  a  useful  reduction 

ia  the  difference  between  the  signal-to-noise  ratios  for  the  high  lights  and  the  low  lights. 

RESOLUTION.    The  resolution  of  the  Iconoscope,  as  for  the  other  pick-up  tubes  here 

eomsidered,  may  be  extended  to  better  than  1000  lines;  for  the  5254ine  standard  it  is 


THE  MONOSCOFE 


15-25 


customary  to  peak  the  high-frequency  response  electrically  so  as  to  keep  the  response  level 
up  to  500  lines. 

UNIFORMITY.  The  description  of  the  operation  given  above  makes  it  evident  that 
the  signal  output  for  any  picture  element  is  not  simply  related  to  its  brightness;  the  re- 
distribution of  the  secondary  and  photoelectrons  makes  it  dependent  both  on  the  geo- 
metrical position  of  the  element  and  on  the  light  distribution  in  the  remainder  of  the 
picture.  Hence  spurious  signals — "shading" — are  introduced  and  must  be  compensated 
electrically  with  the  aid  of  shading  controls.  They  become  particularly  troublesome  when 
large  dark  areas  are  present  in  the  scene. 


11.  THE  MONOSCOPE 

The  monoscope  is  not,  strictly  speaking,  a  pick-up  tube.  It  merely  serves  to  supply  & 
standard  picture  signal,  whose  character  is  prescribed  by  the  preparation  of  the  target 
electrode.  As  such  it  has  found 
application  primarily  in  the 
testing  and  aligning  of  the  com- 
ponents of  a  television  system 
other  than  the  pick-up  tube  it- 
self. 

Figure  9  shows  the  construc- 
tion of  a  type  2F21  monoscope. 
It  consists  of  an  Iconoscope  gun 
and  a  target  plate  (3  Vie  by 
2  5/ie  in.),  normal  to  the  axis  of 
the  gun,  mounted  in  a  pear- 
shaped  envelope.  The  target 


Pattern  electrode 


FIG.  9.     The  Monoscope 


plate  consists  of  sheet  aluminum  on  which  a  pattern  has  been  printed  with  a  carbon  ink 
(Fig.  10).  The  magnetic  beam  deflection  and  the  circuit  connections  are  the  same  as  for 
an  Iconoscope,  with  the  distinction  that  the  anode  coating  is  maintained  at  a  potential 
50-200  volts  positive  with  respect  to  the  target  plate. 


FIG.  10.    Pattern  on  the  Type  2F2I  Monoscope 


The  operation  of  the  monoscope  depends  on  the  much  greater  secondary  emission  ratio 
of  the  slightly  oxidized  aluminum  (~3)  as  compared  with  that  of  carbon  (  <1).    Whenever 


15-26 


ELECTKO-OPTICAL  DEVICES 


the  beam  strikes  the  aluminum,  a  signal  current  approximately  twice  as  great  as  the  beam 
current  flows  through  the  signal  lead  in  such  a  direction  as  to  make  the  grid  of  the  input 
tube  positive;  when  it  strikes  the  carbon,  a  small  current  in  the  opposite  direction  tends 
to  make  it  negative.  If  a  monoscope  replaces  an  Iconoscope  without  changes  in  the  ampli- 
fier, the  portions  of  the  pattern  printed  in  carbon  ink  will  appear  bright  in  the  final  image. 
Dark  and  light  may,  of  course,  be  interchanged  by  adding  or  subtracting  one  stage  in  the 
video  amplifier.  A  peak-to-peak  signal  amplitude  of  several  microamperes  may  be  ob- 
tained with  this  tube. 

12.  THE  ORTHICON 

The  primary  defects  of  the  Iconoscope,  namely,  shading  and  low  efficiency  of  operation, 
both  arise  from  the  redistribution  of  secondary  electrons  and  photoelectrons  on  the 
mosaic.  This,  in  turn,  is  a  consequence  of  the  fact  that  the  equilibrium  potential  of  the 
mosaic  under  the  beam  is  close  to  anode  potential,  in  fact,  slightly  positive  with  respect 
to  it.  If,  however,  the  beam  arriving  at  the  mosaic  has,  initially,  a  kinetic  energy  of  10 
electron  volts  or  less,  the  secondary  emission  ratio  is  less  than  unity  and  the  potential  of 
the  mosaic  will  drop  to  a  value  slightly  below  that  of  the  emitting  cathode;  at  this  equi- 
librium potential  no  additional  electrons  can  reach  the  mosaic.  Instead,  the  beam  elec- 
trons reverse  their  direction  at  a  point  close  to  the  mosaic  and  are  collected  by  some 
electrode  at  positive  potential.  Portions  of  the  mosaic  which  are  in  complete  darkness 
remain  continuously  at  the  equilibrium  potential  and,  hence,  give  rise  to  no  signal  current. 
Illuminated  areas,  on  the  other  hand,  lose  electrons,  between  successive  scannings,  in  exact 
proportion  to  the  quantity  of  light  incident  on  them.  The  low-velocity  beam,  as  it  sweeps 
over  such  areas,  supplies  just  enough  electrons  to  the  mosaic  to  neutralize  the  stored  charge 
and  causes  the  passage  of  an  equal  signal  current  through  the  signal  lead.  In  brief,  the 
actual  operation  of  such  a  low-velocity  Iconoscope  fits  perfectly  the  original,  oversimplified 
version  given  for  the  operation  of  the  Iconoscope. 

The  practical  realization  of  the  low-velocity  Iconoscope  demands  fulfilment  of  two 
conditions:  (1)  in  order  that  the  equilibrium  potential  (and  scanning  spot)  may  be  uniform 
over  the  mosaic,  the  beam  must  be  perpendicular  to  the  mosaic  at  all  points;  (2)  to  keep 
the  effective  spot  size  small,  the  lateral  velocity  components  of  the  electrons  must  be  kept 
small.  These  requirements  are  met  by  the  special  methods  of  beam  deflection  and  focusing 
incorporated  in  the  orthicon. 

The  envelope  of  a  typical  orthicon  (Fig.  11)  is  a  4-in.-diameter  tube  14  in.  long  with  a 
short  neck  for  the  gun  at  one  end  and  a  flat,  clear  window  for  the  transmission  of  the 


Focusing     Anode 
coll  dfsk 

FIG.  11.     Orthicon  with  Electrostatic  Horizontal  Deflection 

optical  image  at  the  other.  It  is  inserted  in  a  long  solenoid  providing  a  uniform  longitudinal 
magnetic  field  of  0.007  weber/m2.  The  envelope  contains,  in  addition  to  the  gun,  a  pair 
of  curved  electrostatic  deflection  plates  which  occupy  half  of  the  tube  nearest  the  gun,  and 
a  mosaic  with  a  translucent  signal  plate  having  a  target  area  2  5/16  by  1  3/4  in.  The  hori- 
zontal deflection  is  accomplished  with  the  aid  of  the  electrostatic  deflection  plates;  the 
vertical  deflection,  by  a  pair  of  coils  mounted  over  a  portion  of  the  second  half  of  the  4-in. 
cylinder. 

The  principal  peculiarities  of  the  deflection  and  focusing  properties  of  the  orthicon  are  a 
consequence  of  the  presence  of  the  longitudinal  magnetic  focusing  field.  In  particular, 
near  the  mosaic,  where  the  magnetic  field  lines  are  perpendicular  throughout  to  the  mosaic 
surface,  the  field  assures  the  normal  incidence  of  the  electron  beam:  In  a  strong  magnetic 
Mid  tlhe  electrons  spiral  about  the  magnetic  field  lines. 

Proceeding  from  one  end  of  the  tube  to  the  other,  the  electrons  leaving  an  indirectly 


THE  IMAGE  ORTHICON  15-27 

heated  cathode  through  an  aperture  in  the  grid  cylinder  (cut-off  voltage,  -40  volts)  are 
accelerated  through  an  aperture  at  225  volts  and  restricted  to  a  narrow  pencil  by  a  defining 
aperture  0.0025  in.  in  diameter  which  is  electrically  connected  to  the  accelerating  aperture. 
This  pencil,  the  scanning  beam,  is  focused  by  the  longitudinal  field,  which  is  strong  enough 
to  form  a  succession  of  images  of  the  denning  aperture  less  than  2  in.  apart.  Its  current  is 
generally  of  the  order  of  0.3  microampere,  i.e.,  just  enough  to  discharge  the  most  strongly 
illuminated  portions  of  the  mosaic.  After  passing  through  another  larger  aperture  in  the 
anode  disk  (at  250  volts),  which  separates  the  gun  chamber  from  the  deflection  chamber, 
the  electrons  enter  the  electrostatic  deflecting  field  between  flared  plates.  The  simultane- 
ous action  of  the  electrostatic  field  and  the  longitudinal  magnetic  field  causes  a  lateral 
displacement  of  the  beam  parallel  to  the  plates  and  proportional  to  the  deflection  voltage 
(160  volts,  peak  to  peak).  The  faring,  making  the  increase  and  decline  of  the  deflecting 
field  gradual,  prevents  the  development  of  cycloidal  loops  in  the  crossed  fields  and,  hence, 
the  acquisition  of  considerable  lateral  velocity  components  by  the  beam  electrons. 

Next,  the  beam  passes  through  the  magnetic  deflecting  field  (0.0025  weber/m2,  peak  to 
peak)  which  simply  warps  the  field  lines,  so  that  the  beam  experiences  a  second  displace- 
ment, in  the  direction  of  the  deflecting  field  (not  at  right  angles  thereto).  After  leaving  this 
deflecting  field  the  electrons  pass  through  a  decelerating  ring  electrode  at  100  volts  (desig- 
nated as  "rotator  electrode/*  since  the  simultaneous  action  of  the  lateral  components  of 
the  decelerating  field  and  the  longitudinal  magnetic  field  causes  a  slight  rotation  of  the 
scanning  pattern)  painted  on  the  envelope  to  the  mosaic,  which  is  inserted  in  a  mask 
maintained  3  volts  negative  with  respect  to  the  cathode.  Both  the  signal  plate  and  the 
photosensitive  mosaic  are  translucent.  Although  the  requirement  of  translucence  reduces 
the  photoemission  of  the  mosaic,  as  compared  with  that  of  the  Iconoscope,  this  is  more 
than  compensated  by  the  greater  efficiency  of  operation. 

SENSITIVITY.  It  is  found  in  practice  that  a  studio  scene  with  a  brightness  of  700 
lumens/m2T  transmitted  with  an  f/2  lens,  will  yield  a  picture  with  a  signal-to-noise  ratio 
of  1 00.  The  target  area  being  0.0026  m2  (4  in.2) ,  the  figure  of  merit,  of  the  orthicon  becomes 


700-0.0026 


2.2  (13) 


This  is  the  same  as  the  figure  for  film  and  better  by  a  factor  of  5  than  that  for  the  Icono- 
scope. However,  since  the  orthicon  does  not  compress  the  brightness  scale  in  the  same 
manner  as  the  Iconoscope,  but  has  a  strictly  linear  response  throughout,  a  signal-to-noise 
ratio  higher  by  a  factor  of  2  or  3  may  be  required  in.  the  transmission  of  naturally  contrasty 
outdoor  scenes  to  attain  an  equal  freedom  from  noise  in  the  low  lights.  Hence,  under 
such  circumstances  a  more  appropriate  value  for  the  figure  of  merit  is  1. 

SIGNAL-TO-NOISE  RATIO  AND  RESOLUTION.  A  signal-to-noise  ratio  of  100  is 
readily  obtained.  Attempts  to  exceed  this  value  by  increasing  the  brightness  of  the  light 
image  frequently  result  in  a  loss  of  resolution  and  a  local  distortion  of  the  scanning  pattern 
at  the  boundaries  between  bright  and  dark  areas. 

UNIFORMITY.  If  the  image  brightness  is  kept  within  the  normal  operating  range  the 
signal  output  yields  a  faithful  representation  of  the  geometry  and  tonal  values  of  the  scene. 
Scene  details  of  excessive  brightness  (e.g.,  the  explosion  of  flash  bulbs)  may,  however, 
cause  a  portion  of  the  scene  to  be  blacked  out.  At  such  points  the  photoemission  charges 
the  mosaic  up  to  a  positive  potential  at  which  the  secondary-emission  ratio  of  the  beam 
electrons  exceeds  unity,  so  that  the  beam  renders  the  illuminated  area  more  positive  instead 
of  discharging  it.  After  the  cause  has  been  removed,  normal  operating  conditions  are 
gradually  re-established  by  surface  leakage. 

13.  THE  IMAGE  ORTHICON 

In  the  image  orthicon  a  very  great  gain  in  sensitivity  has  been  combined  with  the 
freedom  from  spurious  signals  at  low  light  levels  which  is  characteristic  of  the  orthicon 
and  the  stability  of  operation  at  high  light  levels  characteristic  of  the  Iconoscope  at  the 
expense  of  greater  complexity  of  construction  and  alignment.  The  principal  features  which 
distinguish  the  image  orthicon  from  the  ordinary  orthicon  are  (1)  an  electron-optical 
imaging  section,  making  possible  the  employment  of  a  more  sensitive,  continuous,  photo- 
cathode  and  secondary-emission  multiplication  at  the  target;  (2)  a  two-sided  target, 
permitting  limitation  of  the  target  voltage  to  a  value  sufficiently  low  to  insure  stability 
at  all  light  levels  by  providing  a  separate  collector  on  the  side  opposite  to  the  scanned  side; 
and  (3)  a  secondary-emission  signal  multiplier,  for  the  return  beam  current,  which  renders 
the  signal  output  sufficiently  large  that  the  shot  noise  in  the  beam,  rather  than  amplifier 
noise,  determines  the  noise  content  in  the  reproduced  picture. 


15-28 


ELECTRO-OPTIC  Air  DEVICES 


Rorizontaf 
&  vertical 


Photocathode  Target 

\Acceteraior  /Deceterato 
.grid,    / 


Grid 
No.  4 


Alignment 

'"Grid   E'ectro* 
-,--    .,      ,  No.  2    8un 
/    Grid          and      / 
/  No.  3    {  dynode  /  Fire-stage 


Focusing 
c,o]l 


Image 
section 

FIG.  12.     The  Image  Orthieon 


Figure  12  shows,  schematically,  the  construction  of  the  type  2P23  image  orthieon.  The 
tube  has  approximately  the  same  length,  but,  with  the  exception  of  the  short  end  section, 
a  much  smaller  diameter  than  the  1840  orthicon.  As  with  the  latter  tube,  a  long  focusing 
solenoid  must  be  provided  which  envelops  all  the  tube  except  the  gun  and  multiplier 
portion  at  the  right  extremity.  The  light  image  is  projected  on  the  flat  transparent  photo- 
cathode  (maintained  at  —300  volts)  at  the  left  end  of  the  tube,  and  the  photoelectrons 
released  by  the  light  are  focused  by  the  magnetic  field  through  a  very  fine-mesh  (500-1000 
meshes  per  inch),  high-transmission  screen  on  the  target.  Since  the  secondary-emission 

ratio  of  the  glass  target 
screen  for   300-volt   elec- 
trons is  much  greater  than 
unity,   a  positive    charge 
pattern  corresponding  to 
|  NO.  1  /    multiplier    the   light   distribution   in 
\f  /   /    __         the  scene  is  formed  on  the 
target.      The    secondary 
electrons  are  drawn  to  the 
closely  spaced  target 
screen,    which  is   usually 
maintained  at  a  potential 
slightly  positive  (e.g.,  at 
-f-lvolt)  withrespecttothe 
scanning-beam  cathode. 
The  target  itself  is  a  very  thin  disk  of  low-resistivity  glass.    Its  properties  are  such  that 
in  the  course  of  a  frame  time  potential  differences  between  the  two  sides  of  the  target 
built  up  at  the  instant  of  scanning  are  neutralized  by  conduction  while  the  transverse 
leakage  of  stored  charge  between  neighboring  picture  elements  still  remains  negligible. 

The  gun  of  the  image  orthicon  has  the  same  general  construction  as  that  of  the  orthicon. 
However,  the  defining  aperture  is  formed  in  the  final  disk  electrode,  which  is  exposed  to 
the  return  beam.  The  beam  deflection  is  magnetic  throughout,  since  this  causes  the  return 
beam  to  travel  practically  the  same  path  as  the  scanning  beam  in  reverse  direction,  striking 
finally  the  electrode  containing  the  defining  aperture.  The  return  beam  itself  carries  the 
signal;  when  the  smarming  beam  strikes  a  portion  of  the  target  which  has  been  rendered 
positive  by  secondary  emission,  electrons  equal  in  number  to  those  lost  in  the  course  of  a 
frame  time  by  the  element  under  consideration  are  abstracted  from  the  beam,  reducing, 
correspondingly,  the  current  in  the  return  beam. 

The  return  beam  strikes  the  defining-aperture  disk  with  an  energy  of  the  order  of  200 
electron  volts  and  ejects  from  it  secondary  electrons  which,  persuaded  by  the  lower 
potential  of  the  electrodes  facing  it  and  the  higher  potential  of  a  second  stage  of  a  pin- 
wheel  multiplier  structure  surrounding  the  gun,  spill  over  into  the  same  emitting  a 
larger  number  of  electrons,  which  are  drawn  to  the  next  stage.  The  total  gain  of  the  1500- 
volt  five-stage  multiplier  is  from  200  to  500,  which  is  adequate  to  raise  the  shot  noise  level 
in  the  beam  (tV^-10"6  amp)  above  the  amplifier  input  tube  noise  current  (2-10"9  amp, 
both  for  a  5-megacycle  band  width).  At  very  low  lights  the  proper  value  of  the  beam 
current  may,  under  ideal  circumstances,  be  as  low  as  10~"10  amp,  leading  to  a  useful  multi- 
plier gain  of  200;  for  a  high-light  picture  a  gain  of  20  would  suffice. 

SIGNAL  VERSUS  LIGHT  CHARACTERISTICS.  Figure  13  shows  a  typical  variation 
of  the  signal  output  of  the  image  orthicon  with  the  high-light  illumination  of  the  photo- 
cathode.  The  photosensitivity  of  the 
photocathode  is  of  the  order  of  10  micro- 
amperes per  lumen. 

In  the  low-light  range  the  image  orth- 
ieon functions  just  as  the  orthicon,  the 
signal  current  being  proportional  to  the 
light  signal.  At  the  knee  of  the  curve 
the  secondary-emission  charges  the  tar- 
get just  to  the  potential  of  the  target 
screen;  beyond  this  point  the  response 
curve  flattens  out,  since  an  increasing 
number  of  secondary  electrons  are  forced 
to  return  to  the  emitting  element.  As 
the  target  becomes  sufficiently  positive 
to  lose  only  as  many  electrons  by  sec- 


£   J.V 

£ 
T^TO 

See 
Him 

Oe:fc 
tfiha 

tac 

iHon 

<s  and 
salanc 
:'iunj 

wh 
ed 
jstei 

/ 

tes- 

/ 

>tcaf  sflgnaj  oatpt 
o 
OM 

/ 

/ 

2. 

31                  0.1 

1 

10                  10 

£?  Hlghllgnt  illumination  on  photoca'thode,  lux 

FIG.  13.    Signal-versus-Light 'Characteristic  of  Type 
2P23  Image  Orthicon 


ondary  emission  as  it  receives  from  the  photocathode,  the  response  curve  becomes  com- 
pletely, flat.    This  does  not  mean,  however,  than  no  intensity  differences  are  transmitted. 


LUMINESCENT  AND  TENEBRESCENT  MATEKIALS       15-29 

Redistribution  of  secondary  electrons  near  boundaries  between  areas  of  different  in- 
tensity of  bombardment  (different  brightness)  results  in  potential  differences  at  such, 
boundaries.  Thus  a  bright  spot  on  a  less  bright  background  is  transmitted  as  a  bright 
spot  with  a  dark  halo  on  a  bright  background. 

SENSITIVITY.  It  is  found  that  an  image  orthicon  provided  with  an  //2  lens  and 
capable  of  transmitting  a  picture  with  a  signal-to-noise  ratio  of  100  can  do  so  if  the  scene 
brightness  is  20  lumens/in2.  Since  the  target  area  is  approximately  O.QOOS  m2  (1.2  in.*), 
the  figure  of  merit  of  the  image  orthicon  becomes 

92 

=  250  (14) 


20-0.0008 

This  is  approximately  100  times  as  great  as  the  figure  for  film  and  for  the  ordinary  orthicon 
and  over  500  times  as  great  as  that  for  the  Iconoscope.  A  factor  of  5  in  this  gain  must  be 
attributed  to  the  increased  photosensitivity  of  the  photocathode  and  the  secondary- 
emission  amplification  at  the  target;  the  remainder,  to  the  signal  multiplication  in  the 
multiplier.  To  maintain  freedom  from  objectionable  noise  in  the  low  lights  of  high- 
contrast  scenes,  it  may  be  necessary,  just  as  with  the  orthicon,  to  increase  the  signal  by  a 
factor  of  2  or  3,  reducing  the  figure  of  merit  to  unity.  It  may  be  noted  that  the  sensitivity 
of  the  tube  is  sufficient  to  transmit  pictures  with  some  entertainment  value  even  at  a  scene 
brightness  of  0.2  lumen/m2,  corresponding  to  the  brightness  of  light  objects  in  full  moon- 
light. 

RESOLUTION.  The  resolution  of  the  image  orthicon  may  be  limited  by  the  electron- 
optical  imaging  process,  the  target  screen,  transverse  leakage  on  the  glass  target,  and  scan- 
ning spot  size,  the  last  being  influenced  by  the  defining  aperture,  the  angle  of  approach  of 
the  beam  to  the  target,  the  initial  velocity  distribution  of  the  beam  electrons,  and  the 
potential  of  the  scanned  area.  In  practice  it  is  possible  to  attain  a  resolution  of  500  lines 
as  with  the  other  tubes. 

UNIFORMITY.  In  the  low-light  range  (the  sloping  part  of  the  curve  in  Fig.  13),  the 
signal  output  is  a  linear  function  of  brightness.  For  higher  light  values  the  tonal  scale  is 
compressed,  and,  ultimately,  contrasts,  rather  than  absolute  light  values,  are  transmitted 
primarily.  This  condition  does  not  detract  materially,  however,  from  the  apparent 
naturalness  of  most  reproduced  pictures.  Geometric  distortions  are  inappreciable,  al- 
though slight  non-uniformities  in  the  target  and  the  presence  of  the  target  screen  tend  to 
make  the  picture  somewhat  inferior  to  that  transmitted  by  the  other  pick-up  tubes  de- 
scribed. 

14.  FIELDS  OF  APPLICATION  OF  PICK-TJP  TUBES 

The  characteristics  of  the  several  pick-up  tubes  here  discussed  mark  out  spheres  of 
application  for  which  each  is  particularly  suitable.  Thus,  the  relatively  insensitive  image 
dissector,  with  its  freedom  from  signal  distortion,  may  be  employed  for  the  transmission 
of  motion  pictures,  for  which  very  high  light  levels  can  readily  be  provided.  The  standard 
Iconoscope  is  well  suited  for  both  movie  and  studio  work,  where  the  light  distribution  can 
be  controlled  so  as  to  simplify  the  compensation  of  shading.  Its  smaller,  2-in.  version  is  a 
convenient  television  pick-up  device  for  industrial  and  experimental  purposes.  Spot  pick- 
up, with  the  attendant  unpredictable  conditions  of  lighting,  demands  the  employment  of 
the  image  orthicon  which,  in  view  of  its  greater  complexity  and  somewhat  inferior  picture 
quality,  may  under  other  circumstances  be  replaced  advantageously  by  the  less  sensitive 
tubes.  It  is  to  be  hoped  and  expected  that  further  development  of  the  image  orthieon,  as 
the  most  recent  of  the  pick-up  devices,  will  raise  the  level  of  its  picture  quality  to  that  of 
the  older  pick-up  tubes. 


LUMINESCENT  AND  TENEBRESCENT  MATERIALS 

By  H.  W.  Leverenz 

Luminescence  is  a  production  of  light  in  excess  of  thermal  radiation  (see  ref .  5  on  p.  15-41) . 
Thermal  radiation  is  emitted  by  electrons,  atoms,  ions,  and  molecules  oscillating  or 
rotating  singly  or  in  groups  as  occasioned  by  thermal  agitation. 

An  ideal  thermal  radiator  is  the  perfect  ttack  body,  which  has  complete  absorptivity  at 
all  wavelengths;  i.e.,  it  has  oscillators  available  at  all  frequencies.  The  monochromatic 
emissive  power,  EVr  of  a  perfect  black  body  (in  vacuum)  at  frequency  v  (in  sec"1),  is  a 


15-30 


ELECTRO-OPTICAL  DEVICES 


ir 


si 
si 


>s 
H 


EXCITATION    SPECTRUM 


•  SILICATES  (ALSO 
PICAL  OF  TONCSTATf 
tD    BORATES) 

A 

/\ 


I 
I 
t 

1 
I 
1 
I 

\ 


EMISSION    SPECTRA 
OF   TYPICAL  SILICATE 


GftEEN  LUMINESCENCE) 


function  of  temperature,  T  (In  degrees  Kelvin),  according  to  Planck's  radiation  law: 

Ev  =  4.63  X  icr5V(e7lJ'>'*T  -  I)"1    watts/m2  (1) 

where  h  =  6.624  X  10  ~34  joule-sec  (Planck's  constant). 
8  =  2.71828*  -  -  (base  of  Napierian  logarithms). 
k  =  1.38  X  IQ-^Joule/deg  (Boltzmann's  constant). 

The  peak  wavelength,  Xmax,  of  the  broad  emission  band  of  black-body  radiation  varies 
with  absolute  temperature  according  to  Wien's  displacement  law: 

Xmax  =  2.897  X  IQ^T'1  m  (1  m  =  105  microns  (/t)  =  1010  angstrom  units  (A))      (2) 

The  total  emissive  power,  ET,  Ma  black  body  is  proportional  to  the  fourth  power  of  the 
absolute  temperature  according  to  the  Stefan-Boltzmann  law: 

E*r  -  5.67  X  lO-8^4    watts/m2  (3) 

At  room  temperature,  «300  deg  Kelvin,  Xmas  is  in  the  far  infrared  at  9.7  microns,  and  ET 
is  only  459  watts/m2.  It  should  be  noted  that  the  nature  of  the  material  plays  no  role 
in  eqs.  (1),  (2),  and  (3).  Although  thermal  radiation  is  emitted  by  all  materials  at  tem- 

peratures greater  than  0  deg  Kelvin 
such  radiation  does  not  become  visible 
until  the  temperature  is  raised  above 
about  1000  deg  Kelvin,  when  incan- 
descence is  observed.  A  temperature 
of  the  order  of  6500  deg  Kelvin  is  re- 
quired to  shift  the  peak  wavelength, 
Xmax  of  thermal  radiation  into  the 
visible  region  of  the  spectrum  £4000  to 
7000  1).  Thus  far,  no  solid  material 
has  been  developed  to  endure  pro- 
longed operation  above  4000  deg  Kel- 
vin, and  so  most  of  the  energy  emitted 
from  solid  incandescent  materials  lies 
in  the  infrared  and  their  efficiencies  of 
light  production  are  generally  less  than 
about  5  per  cent. 

Luminescence  is  occasioned,  by  ab- 
sorbed photons,  so-called  undulatory 
energy  (e.g.,  ultraviolet,  x-rays,  7- 
rays),  or  corpuscular  energy  (e.g., 
cathode  rays  or  a-particles)  which  ex- 
cite electronic  transitions  directly 
rather  than  through  the  intermediate 
stage  of  thermal  agitation  of  atoms 
and  ions.  Luminescence  emission  is 
usually  in  the  form  of  spectral  lines  or 
narrow  bands  superimposed  on  the 
broad  band  of  thermal  radiation  from 
a  material.  The  spectral  distributions 
and  efficiencies  of  luminescent  mate- 
rials  are  determined  largely  by  their 
chemical  compositions  and,  if  the  ma- 
terials are  solids,  by  their  crystalline 
structures.  The  characteristic  mono- 
chromatic spectra  of  attenuated  gases 


4ZitO-6BtO'3SiOx:Mn 
(YELLOW-ORANGE       • 
LUMINESCENCE) 


3000 

ULTRAVfOLET- 


5000  6000 

-  VISIBLE 


XX)  7000A* 


EMISSION    SPECTRA 
OF  TYPICAL   SULPHIDE 
r,  PHOSPHORS 


200O 

PIG.   1. 


j5 


Excitation  and   Emission  Spectra  of  Some 
Typical  Phosphors 


are  relatively  simple  luminescences  whose  efficiencies  may  approach  100  per  cent  for  the 
case  of  resonance  radiation  (Xexcltation  =  Demission)-  In  liquids  and  solids,  however,  the 
perturbations  imposed  by  near  neighbors  of  a  luminescing  atom  or  ion  complicate  the 
mechanism  and  generally  lower  the  efficiency  of  luminescence. 

The  generic  term  luminescence  is  commonly  modified  by  a  prefix  indicative  of  the 
excitant  used  to  cause  luminescence.  For  example,  photoluminescence  is  luminescence 
excited  by  photons,  and  cathodoluminescence  is  luminescence  excited  by  cathode  rays.  A 
further  distinction  is  made  with  respect  to  duration  of  luminescence  after  cessation  of 
excitation;  i.e.t  fluorescence  lasts  less  than  about  10~8  second  whereas  ^phosphorescence  lasts 
longer  than  about  10  ~8  second.  The  value  of  10  ~8  second  is  the  approximate  lifetime  of 
eaceited  non-metastable  isolated  atoms  or  ions  and  serves  as  an  arbitrary  demarkation 
between  fluorescence  and  phosphorescence.  Materials,  such  as  gases,  liquids,  organic 
materials,  and  many  glasses  which  exhibit  fluorescence  are  called  flitors;  while  phosphores- 


LUMINESCENT  AND  TENEBRESCENT  MATERIALS       15-31 


CURVE 

PHOSPHOR 

CRYSTALLIZATION 
TEMPERATURE  *C 

EXCITATION 

, 

<x*-ZnS    IO"*Ni 

73O 

3650 

*»-Z«S   0.003%  Aq 
P*-ZnS    0.015  %Aq 
a*-Z«S    O.OO3%CM 

950 
1240 
660 

3660 
3*50 
36iO 

/3*-Z«S   O.OO31&OJ 

\20Q 

3650 

j3*-Z«S   O 

1260 

3650 

1200 

2537 

{3  -  Znj  Si  CU  :  I  fc  Mn 

I55O-O 

2537 

ZnO:CZ«0 

IOOO 

365O 

300     J 
600 


TEMPERATUPE 


PHOSPHORESCENCE  • 


F- SHORT- PERSISTENCE    PHOSPHOR 
P  =  LONG  -  '*  ii 


FIG.  2.     Photoluminescences  of  Some  Phosphors  as  a  Function  of  Temperature 

cent  materials,  which  are  chiefly  crystalline  inorganic  materials,  are  called  phosphors. 
Some  typical  excitation  and  emission  spectra  of  phosphors  are  shown  in  Fig.  1;  several 
temperature-dependence  curves  of 
phosphor  photoluminescences  are 
shown  in  Fig.  2,  and  some  typical 
excitation  and  decay  characteris- 
tics of  phosphors  are  indicated  in 
Fig.  3. 

Phosphor  light   outputs  may  be     I 
modulated  by  three  methods:  & 

1.  Positive  modulation  of  lumines-     5 
cence  is  the  normal  increase  of  light     « 
output    with    increasing    excitation     ? 
density  at  temperatures  below  the     | 
fairly  critical  temperature,  Tc,  above     t 
which  the  efficiency  of  luminescence     ° 
sharply  decreases.  t 

2.  Negative  modulation  of  lumi-     | 
nescence  is  accomplished  by  increas-    £ 
ing  the  temperature  of  an  excited    '" 
phosphor  above  Te  and  thereby  de- 
creasing the  luminescence. 

3.  Positive  modulation  of  incan- 
descence is  accomplished  by  further 
raising  the  temperature  of  the  phos-      < 
phor  until  incandescence  supplants    p^  ^ 
luminescence. 


EXCITATION 
INTERVAL  "" 


DECAY  INTERVAL  - 


The  Relationships  of  Luminescence,  Fluorescence, 
and  Phosphoresceace 


15-32 


ELECTRO-OPTICAL  DEVICES 


The  last  two  methods  of  modulation  involve  thermal  inertia  of  matter  as  contrasted 
with  the  purely  electronic  transitions  in  positive  modulation  of  luminescence.  Positive 
modulation  of  luminescence  is  unique  in  allowing  useful  modulation  up  to  frequencies  of 
the  order  of  107  cycles  per  second,  the  limit  for  any  particular  phosphor  being  inversely 
proportional  to  its  characteristic  decay  time  (the  time  taken  to  decay  to  an  arbitrary 
percentage,  e.g.,  1  per  cent,  of  the  luminescence  at  the  last  instant  of  excitation). 

Tenebrescence  is  any  non-intrinsic  absorption  of  light  induced  in  a  material.     For 

example,  normally  colorless  potassium  chloride,  KC1,  whose  intrinsic  absorption  is  in  the 

^  rt0       o  ,,-r     «  ne     ..  T-7  M     far  ultraviolet  (left  side  of  Fig.  4),  may 

12.34      6.17        4.11       3.08         2.47       2£6      1.77  *V  * 

JOOO       2000      3000      4000       5000       6000     7000  A        ^.^  J^  spectrum  by  irradia- 

tion  with  cathode  rays  or  x-rays  (see 
right  side  of  Fig.  4).  The  induced  dark- 
ening (tenebrescence)  may  be  bleached 
by  irradiating  the  darkened  material 
with  light  having  wavelengths  lying 
within  the  induced  absorption  band. 
Intrinsic  and  Induced  (Tenebrescence)  Absorp-  The  Bleaching  is  accelerated  by  heat. 


Intrinsic 


Induced 


FIG.  4. 


tion  Bands  of  Potassium.  Chloride 


Tenebrescent  materials  become  increas- 


ingly difficult  to  bleach  as  the  duration  and  intensity  of  the  primary  irradiations  used  to 
induce  tenebrescence  are  increased.  The  relatively  unbleachable  absorptions  are  similar 
to  those  of  pigments  or  dyes  which  convert  absorbed  photons  into  heat.  Bleachable 
tenebrescences  are  ascribed  to  temporary  trapping  of  electrons;  unbleachable  tenebres- 
cences  apparently  involve  concomitant  ionic  displacements. 

Tenebrescent  materials,  such  as  the  crystalline  halides  of  alkali  or  alkaline-earth  metals, 
are  called  scotophors. 


15.  PREPARATION  AND  NOTATION  OF  PHOSPHORS 

Successful  preparations  of  synthetic  phosphors  require  highly  specialized  chemical  and 
physical  operations  wherein  even  the  most  skilled  and  careful  workers  sometimes  have 
difficulty  in  reproducing  results.  Phosphor  ingredients  must  be  purified  to  contain  less 


CURVE 

PHOSPHOR 

RELATIVE  VISIBLE 
EFFICIENCY 

COLOR 

; 

a-Zi>»SiO4:Mn(STCO 
tf-Z**&CV-Mn 

100 
60 

GREEN  (5230  A) 
YELLOW  (563O  £) 

3000 
UUWS/tOLET 


4OOO 
VIOLET 

WAVELENGTH  -ANGSTROMS 


*  5.    Cathodolumlnescence  Spectra  of  «-  and  ^-Z 


with  and  without  Manganese  Activator 


than  about  10~*  per  cent  of  undesirable  metallic-ion  impurities  (e.g.,  iron,  nickel,  and 
eliromiiim),  since  as  little  as  10~*  per  cent  of  combined  nickel  in  a  zinc-cadmium-sulfide 
pi*ospfeor  lowers  efficiency  about  25  per  cent.  On  the  other  hand,  10  ~*  per  cent  of  com- 
bined silver  in  the  foregoing  pure  phosphor  increases  efficiency  100  per  cent.  The  final 
step  in  preparing  phosphors  is  crystallization,  where  the  purified  ingredients  are 


PREPARATION  AND  NOTATION  OF  PHOSPHORS      15-33 


CURVE 

PHOSPHOR 

TEMP. 
FOR 

RELATIVE 
VISUAL 

RESPONSE 

NATURAL 
COLOR 

COLOR  OT 
LUMINESCENCE 

RELATIVE  ENERGY  RELATIVE  VISUAL  RESPONSE 

.  §  8  §  8  §  o  §  8 

1 

a 

3 

5 
6 
7 

ZnS:O.O08%A9 

940° 

»8.5(la) 

WHITE 

LIGHT    BLUE 

ZnS(80)  CdS(20)'O.OI%Ag 

'• 

2T.O 

LIGHT 
GREEN    WHITE 

VERY    LIGHT 
BLUE   GREEN 

ZnS(60)ClS(40):        " 

66.3 

VERY 
LIGHT   GREEN 

VERY    LIGHT 
CREAM    GREEN 

Z*S(50)CdS(50):       •« 

" 

IO&OG-) 

LJGHT    YELLOW 

LIGHT    GREEN 
YELLOW 

Zr.S(-30)CdS(60):       » 

" 

63.7 

LIGHT 
CREAM    YELLOW 

LIGHT 
YELLOW  ORANGE 

ZnS(20)CdS{60):        •• 

•  • 

9.4(6a) 

TAN  ORANGE 

LIGHT    RED 

0,5:0.02%*, 

» 

— 

LIGHT 
BROWN    ORANGE 

RED 

1 

/ 

IN 

a 

S^~ 

^ 

IEYE 

[MAX. 

^ 

*c 

» 

vioirr 

5 
BLUC 

00 

CBCCM 

M 

rtttow 

00 

OftAMCE 

TO 
KO 

•00 

1 
I 

f 

/ 

\ 

'EYE 

I  MAX. 

I 

/ 

V 

^\ 

I 

/ 

fr 

X/^ 

P" 

\    > 

^ 

u, 

v  '/ 

^ 

IVJ.x 

\ 

&- 

5000  600O 

WAVELENGTH  -ANCSTROM    UNITS 


PIG.   6.    Cathodoluminescence    Emission   Spectra   of   Some    Silver-activated    Zinc-cadmium-sulfide 
Phosphors.    Upper  curves  are  the  relative  visual  response  characteristics  of  Nos.  1,  4,  and  6. 


Violet 


Bftfe 


Green 


Yeftow       Orange 


mixed  in  fused-silica  or  platinum  crucibles  and  heated  in  electric  resistance  furnaces, 
generally  to  temperatures  be- 
tween 600  and  1600  deg  cent. 
The  resultant  phosphors  are 
masses  of  tiny  crystals  ranging  400 
from  less  than  0.01  to  about  100 
microns  in  diameter.  Most  phos- 
jphor  crystals  average  about  1  to 
15  microns  in  diameter.  Some 
-iypical  initial  compositions  and 
t  corresponding  notations  of  the 
resultant  phosphors  are  given  in 
•Table  1. 

The  luminescence  emission 
:  spectra*  of  phosphors  are  strongly 
.influenced  by  changes  in  crystal- 
Ilization  and  composition,  as 
.  shown  in  Figs.  5,  6,  and  7.  Phos- 
.phors  such  as  P3,  P4(Y),  and 
P7/2  belong  to  "families" 
wherein  gradual  base-material 
-variations  enable  one  to  produce 
emission  spectra  which  may  be 

-  varied  continuously  from  one  end 
«of  the  visible  spectrum  to  the 

*  other.    Other  properties,  such  as 


4500 


5000  5500 

Waffielengtb,  angstrom  unfts 


6000 


65OO 


FIG.  7.     Cathodoluminescenoe  Emission  Spectra  of  Zinc-sul- 

fide  Phosphors  Prepared  with  (1)  no  added  activator,  (2)  copper 

activator,  (3)  silver  activator,  and  (4)  gold  activator 


15-34 


ELECTRO-OPTICAL  DEVICES 


absorption  spectrum,  efficiency,  and  phosphorescence,  are  also  considerably  affected  by 
changes  in  the  structures  and  compositions  of  phosphors. 

Table  1.     Approximate  Compositions  and  Notations  of  Some  Useful  Phosphors 


RMA 
Code 

Base  Material 
Ingredients, 
grams 

Activator 
Salt,  grains 

Flux, 
grams 

Crystal- 
lization 
Temper- 
ature, 
deg  cent 

Phosphor  Notation 

PI  
P2 

P3 

P4(Y) 
P4(B) 
P6(B) 
Pll 
P4(Y) 
P5 
P6(G) 
P6(R> 
P7/1 
PI4/1 
P7/2 
P12 
PI  4/2 
P15 

81  ZnO  -f-  31  SiO2  
IDOZnS 

0.5  MnO 
0.02CuCl2  + 
0.04  AgCl  t 
0.4  MnO 

0.002  to  0.02 
AgCl 

0.02  AgCl 

1250 
1250 

1250 
950 

950 
1000 
950 
950 
1250 

1250 
1000 
1200 

1000 

tf-Zn2SiO4:Mn 
0*-ZnS:Ag  t:Cu 

8ZnO  -BeO  -  SSiOa  :  Mn 
a  *-ZnS:Ag 

ZnS(48)-CdS:Ag 
CaW04:[W] 
ZnS(60)-CdS:Ag 
ZnS(38)-CdS:Ag 
(3  *-ZnS:Ag 

Zn?(86)-CdS:Cu 
ZrJFo:Mn 
ZnS(75)-CdS:Cu 
ZnO:[Zn] 

SrS  :  SrSe  :  Sm  (Tb)  :  Eu  (Ce  ) 

6NaCl 

65  ZnO  4-  2.5  BeO  + 
31  Si02 
100  ZnS     

2NaCl 
2  NaCl 

48  ZnS  +  52  CdS 

57  CaO  4-  232  WO3  .  . 

60  ZnS  -f  40  CdS 

0.02  AgCl 
0.02  AgCl 
0.02  AgCl 

0.02  CuCl* 
0.5  MnF2 
0.01  CuCl2 
(Heat  in  CO 
1000° 
0.03  SmCls  (or 
TbCla)  4- 
0.03 

Eu-2(SO4)3 

(or  CeCis) 

2NaCl 
2NaCl 
4  NaCl  + 
2  BaCl2 
2  NaCl 

38  ZnS  +  62  CdS     .    . 

100  ZnS 

86  ZnS  4-  14  CdS.    ..  . 
103  ZnFa      .  .  . 

75  ZnS  4-  25  CdS 

2  NaCl 
or  H2  at 
C) 
6  (CaF2  4- 
SrS03) 

100  ZnO   ... 

100  SrS  (or  SrS  + 
SrSe) 

a  *  =  cubic;  #  *  =  hexagonal. 
f  =  optional.     (B)  =  blue,    (G)  =  green,   (Y) 
ions  may  also  be  added  as  nitrates. 


yellow,   (R)  =  red.     The  Ag  and   Cu   activator 


16.  MECHANISMS  OF  PHOSPHORS 

Energy  transducfions  during  excitation  and  emission  of  phosphor  luminescences  have 
the  following  chronological  sequence: 


—  03)  ^internal  fluorescence  ~}~ 


-^primary  =  ^reflected  +  ^^absorbed  +  cJ^escaped 

—  ^i-Stransmltted  to  activator  centers  Hr  (&  —  &i)  •S'lieat 
activator  centers  =  ^2-^stored  ~f"  (&i  —  ^2)  ^Internal  fluorescence 
phosphorescence  +  fa 
orescence  — 
luminescence  ~h  ( 


(5  l 


where  a  +  b  +  c  =  1;  1  >  b  >  bi  >  h*  >  h$  >  h:  and  -^escaped  is  the  residual  primary 
energy  which  completely  penetrates  the  phosphor  crystals  or  which  emerges  from  the 
side  of  incidence  owing  to  internal  scattering. 

Luminescence  emission  is  occasioned  when  a  bound  electron  in  energy  state  E$  is  excited 
to  a  higher  allowed  energy  state,  #ex»  and  returns  to  the  same  or  an  intermediate,  energy 
level,  E&ct,  emitting  the  energy  difference,  AJ?  =  J5?ex  —  E&r,t,  as  a  photon  of  light.  The 
relations  between  energy  AJ?  (in  joules),  frequency  v  (in  cycles  per  second),  and  wave- 
length X  (in  meters),  of  photons  are  given  by: 


AE  =  hv  = 


(9) 


where  c  =  3  X  10s  m/sec  (speed  of  light  in  vacuum). 

Some  of  the  features  of  corpuscular  excitation  of  phosphors,  such  as  by  cathode  raysr 
may  be  exemplified  with  the  aid  of  Fig.  S,  which  shows  generalized  sketches  of  the  interiors 
ol  pbosphor  crystals,  including  three  major  classes  of  crystal  irregularities  (faults).  The 
total  penetraiionT  a?fT  of  10s  to  106  volt  cathode  rays  in  a  phosphor  of  density  a  (in  grams 


MECHANISMS  OP  PHOSPHORS 


15-35 


per  cubic  centimeter)  is  calculable  from  Ten-ill's  equation: 


4  X  10ncr 


cm     (To  in  volt  si 


(10) 


The  fraction  W/WQ,  representing  the  power  dissipated  up  to  distance  x  in  the  crystal,  is 
given  by  StinchfiekTs  equation: 


W  /          x 

=r  -  1  -  -y(  1 

WQ  \  xt 


Cfx/xt 


Charged-partide 

excitation 


(11) 

where  7  «  1  for  x/xt  <  0.5  (7  increasingly  exceeds  unity  for  x/'xt  >  0.5) ,  and  C'  ~  32. 
Over  50  per  cent  of  the  cathode-ray  power  is  dissipated  in  the  first  quarter  of  the  total 
penetration  distance,  and  over  80  per  cent  is  dissipated  in  the  first  half  of  the  total  pene- 
tration distance. 

A  few  phosphors,  such  as  ZnO:[Zn],  may  be  excited  by  cathode  rays  with  energies  as 
low  as  5  volts,  but  such  low- voltage  excitation  is  quite  inefficient  because  the  excitation 
energy  is  expended  in  the  distorted 
surface  layers  of  the  phosphor  crys- 
tals and  the  ratio  of  secondary  to 
primary  electrons  is  usually  less 
than  unity  at  such  low  primary" 
voltages.  Conventional,  unmetal- 
lized  cathode-ray-tube  screens  must 
have  secondary-emission  ratios  equal 
to  or  greater  than  unity  to  main- 
tain .a  positive  potential  with  re- 
spect to  the  cathode.  The  efficient 
range  of  primary  voltages  is  above 
about  1000  volts,  and  preferably 
above  10,000  volts  for  phosphors 
whose  limiting  potentials  (voltage 
above  which  the  secondary-emission 
ratio  falls  below  unity)  are  above 
10,000  volts  or  phosphor  screens 
which  may  be  coated  with  an  elec- 
tron-pervious reflecting  and  conduct- 
ing coating,  such  as  a  1000-A-thick 
layer  of  aluminum. 

As  indicated  in  Fig.  8,  the  initial 
energy,  eV&  of  the  primary  particle 


II 

I  *',  h"z 


a  =  substitutions!  fmpurrty 
&  "interstitial  Impurity 
O  =  omission  defect 


("quantum  energy  emitted-}-  (heat  dev.) 
y<  \heat  developed 

I  escape  energy  4-  heat  dev. 


fn 

f absorbed ^ 

"<  In 


is  expended  bitwise  and  indiscrimi-   ^  JttwMp*tion\>f * P 

nately  to  the  crystal  atoms  and  ions.  Single  Absorption  of  a  Primary  Photon 

The  sizes   of  the   absorbed  energy 

bits  average  about  20  to  30  electron  volts,  as  determined  by  the  characteristic  frequencies 
of  bound  electrons  in  the  crystal  (1  electron  volt  =  1.6  X  1Q~19  joule  =  1.6  X  10 ~^  erg). 
The  average  absorbed  energy  bits  are  relatively  independent  of  the  nature  of  the  inorganic 
material  and  the  initial  energy  of  the  primary  particle.  A  high  degree  of  crystallinity  is 
essential  for  efficient  cathodoluminescence,  since  the  indiscriminately  absorbed  energy 
bits  must  be  transmitted  to  the  sparse  population  of  activator  (phosphorogen)  centers 
with  the  minimum  of  attenuation.  Glassy  structure,  crystal  faults,  and  undesirable 
impurities  lower  luminescence  efficiency  by  converting  absorbed  primary  energy  into  heat 
as  indicated  in  eqs.  (5)  and  (7).  The  efficiency  loss  (as  heat)  in  eq.  (8)  is  occasioned  by 
the  absorptivity  of  the  phosphor  crystal  for  its  own  luminescence.  The  more  efficient 
cathodoluminescent  materials  require  an  average  of  over  30  electron  volts  of  primary 
cathode-ray  energy  per  1.5  to  3  electron  volt  quantum  of  emitted  luminescence.  Hence, 
the  efficiency  of  cathodoluminescent  materials  has  thus  far  been  less  than  about  10  per 
cent. 

In  photon  excitation  of  phosphors,  the  primary  photon,  hvo,  seeks  out  a  spot  in  the  crystal 
to  expend  itself  completely,  not  bitwise.  The  absorption  of  a  primary  beam,  containing 
no  photons  per  unit  cross-sectional  area,  is  a  function  of  penetration  distance,  xt  into  the 
phosphor,  according  to:  _  _As  .  . 

where  the  absorption  coefficient  A  is  strongly  dependent  on  the  frequency  of  the  primary 
photon  and  the  characteristic  allowed  frequencies  in  the  phosphor  crystal.  The  char- 
acteristic frequencies  of  bound  electrons  in  the  base  materials  comprising  the  bulk0of 
phosphor  crystals  lie  in  the  far  ultraviolet  (v  >  15  X  1014  cycles  per  second,  X  <  2000  A) 


15-36  ELECTRO-OPTICAL  DEVICES 

Excitation  of  phosphors  by  near-ultraviolet  photons  results  in  direct  excitation  of  the 
foreign  activator  centers,  and  there  is  no  need  for  a  high  degree  of  crystallinity.  Good 
photoluminescence,  but  inefficient  cathodoluminescence,  is  obtained  from  amorphous 
materials  such  as  organic  dyes,  inorganic  glasses,  and  certain  inorganic  crystals  such  as 
the  alkaline-earth-sulfide  phosphors  which  contain  some  glassy  structure  caused  by  the 
residual  non-volatile  fluxes  used  in  their  preparation. 

Photoluminescence  efficiency  is  limited  both  by  impurities  and  crystal  faults,  which 
convert  primary  photons  and  internal  luminescence  photons  into  heat  according  to  eqs. 
(5),  (7),  and  (8)r  and  by  the  energy  deficit,  A#: 


-  Emitted)  ~ 

Efficient  phosphors,  excited  by  near-ultraviolet  photons,  have  photolumin  escence  efficien- 
cies of  the  order  of  60  to  SO  per  cent,  with  quantum  efficiencies  near  unity.  At  very  high 
energies  of  primary  photons  (or  particles)  the  production  of  secondary  radiations  intro- 
duces added  complications  in  the  mechanisms  of  excitation  of  phosphors. 

There  are  two  major  types  of  luminescence-active  centers  in  phosphor  cyrstals: 

Stibstitutionally  located  activators  are  exemplified  by  manganese  which  replaces  zinc, 

especially  in  phosphor  crystals  where  oxygen  or  fluorine  dominate  the  anion  structure  (e.g., 

zinc  silicate,  and  zinc  fluoride).     These  phosphors  afford  predominantly  simple  initial 

exponential  (&~af)  decays  of  light  output,  L,  after  excitation  to  a  peak  luminance,  LQ,  accord- 

ing to: 

L  =  LQQ-at        (t  in  seconds)  (14) 

where  the  constant  a  has  known  values  ranging  from  106  for  ZnO:[Zn]  to  10  for 
(Zn:  Mg)F2.*  Mn.  Exponential  decays  are  determined  largely  by  the  chemical  composition 
of  the  phosphor,  being  relatively  unaffected  by  changes  in  crystal  structure,  in  tempera- 
ture, or  in  the  type,  duration,  or  intensity  of  excitation.  The  luminescence  action  is 
apparently  localized  in  the  substitutionally  located  activator  sites  as  a  metastable-state 
monomolecular  process,  without  necessitating  electron  transport  outside  the  immediate 
sphere  of  influence  of  the  substitutional  activator  center.  The  observed  weak  photo- 
conductivities of  some  8~af-decay  phosphors  are  probably  associated  with  their  later-stage 
low-intensity  power-law-decay  "tails"  which  are  strongly  affected  by  changes  in  crystal 
structure,  temperature,  and  excitation. 

InterstitiaUy  located  activators  are  exemplified  by  copper  or  silver  interspersed  among 
the  regular  lattice  units  in  phosphors  where  sulfur  and/or  selenium  dominate  the  anion 
structure.  These  phosphors  afford  so-called  power-law  (t~n]  decays  represented  by 


where  both  6  and  n  are  not  true  constants  but  vary  with  changes  in  £0,  t,  crystal  structure, 
or  temperature,  and  with  the  type  and  duration  of  excitation.  For  practical  purposes, 
0  <  b  <  10  ~3  and  0.2  <  n  <  3  for  known  phosphors.  There  is  a  definite  correspondence 
between  the  strong  photoconductivities  and  phosphorescences  of  most  £~n-decay  phos- 
phors, indicating  electron  transport  between  the  remote  centers  of  trapping  and  emission. 
In  these  cases,  monomoleeular  activated  release  followed  by  bi-  or  polymolecular  mech- 
anisms afford  the  complex  £""*  decays. 

The  optimum  activator  concentrations  and  maximum  allowable  impurity  concentra- 
tions in  e~~ct-decay  phosphors  are  about  a  hundredfold  greater  than  those  in  £~n-decay 
phosphors.  Since  the  cube  root  of  100  is  about  5,  this  means  that  interstitially  located 
impurities  are  affected  by  other  impurities  five  times  as  remote,  as  in  the  case  of  substitu- 
tionally located  impurities.  Substitutionally  located  impurities  are  in  regions  of  lower 
potential  energy  and  are  better  buffered  by  close  coupling  with  the  lattice  forces. 

The  optimum  number  of  active  luminescence  centers  and/or  electron  traps  is  about  1017 
per  cubic  centimeter  in  £~n-decay  phosphors  and  about  1019  per  cubic  centimeter  in  8~  at- 
decay  phosphors.  From  these  data,  and  data  on  the  penetrations  and  power  densities  of 
common  excitants,  it  is  possible  to  calculate  the  maximum  phosphorescences  of  phosphors 
with  given  decays. 

17.  MECHANISMS  OF  SCOTOPHORS 

Figure  9  shows  a  schematic  section  of  a  crystal  of  potassium  chloride,  KC1,  indicating 
several  omission  defects  and  an  interstitial  defect.  The  fraction  /  of  omission  defects  is  a 
function  of  equilibrium  temperature  T  (degrees  Kelvin)  according  to 


wfeare,  for  KC1,  A  «  104  and  TF0  »  1.6  X  10~19  joule.    At  temperatures  near  the  melting 


SPECIFIC   CHARACTERISTICS  OF- USEFUL  PHOSPHORS      15-37 

point  of  KCI  (1074  deg  Kelvin),  /  becomes  about  0.2,  and  many  of  the  omission  defects 
are  frozen-in  when  KC1  is  evaporated  and  condensed,  as  in  making  P10  screens.  There- 
fore, a  large  number  of  the  3.2  X  1022  lattice  sites  per  cubic  centimeter  in  PlO-screen 
crystals  are  empty,  and  half  of  these  omission  defects  are  absent  chlorine  ions  (Cl~)  whose 
vacant  positions  may  be  occupied  by  electrons  to  form  scotophor  color  centers  (so-called 
F-centers)  as  indicated  in  Fig.  9.  The  visible  absorption  band  of  tenebrescent  KC1  (Fig.  4) 
corresponds  to  absorption  of  light  by  F-centers,  i.e.,  ejection  of  the  trapped  electrons. 
The  maximum  concentration  of  F-centers  in  KC1  screens  is  about  101S  F-centers  per  cubic 
centimeter. 

Tenebrescence  is  quantitatively  expressed  as  contrast,  C,  defined  by 

c_  1000. -i«>      (inpercent)  (17) 

LQ 

where  I/o  and  Ld  are  the  luminances  of  the  undarkened  and  darkened  areas,  respectively, 
when  observed  under  light  having  wave-  _  _  /  _ 

lengths  within  the  induced  absorption  band  of  O    O*O    O*         P+O 

the  scotophor.  -._        ,-_ 

The  decay  of  tenebreseence,  i.e.,  the  bleach-  U  P   U   O    ( 

ing  of  induced  darkening,  is  a  power-law  rela-  rv*rT 

tion  of  the  general  type  u   ^ 

e  =  eo*-»  as) 

where  C0  is  the  tenebrescence  (contrast)  at      O*  O~  O*  tfj?  O4  Q~  O+  O"" 

time  t  —  0  and  n  decreases  from  about  1.5  to      ^_        /-v_     ,  s-\-*+  s~\-  ^3.  /~\-  r*-J- 
less  than  0.1  with  decreasing  temperature  or      U    °   (J   O*Vj^O   {J    VT^J    U 
intensity  of  illumination  and  with  increasing  CR-beam  electron, 

degree  of  tenebrescence.     The  useful  decay      FIG.  9.     Schematic  Section  of  a  Crystal  of  Po- 
intervals    (time   between   successive   excita-     tassima  Chloride  (KCI),  showing  the  formation 
tions)    of  the   best   known  scotophor,   KCI,     of  an  F-center-  ^S^^.  "  C1~r  ""^ 
range  between  1  and  60  seconds  at  tempera- 
tures near  40  deg  cent  and  illuminations  of  the  order  of  10,000  foot-candles  of  light  from 
incandescent  lamps. 

The  sensitivity  of  cathodotenebrescent  KCI  is  about  1/ioo  that  of  efficient  cathodo- 
luminescent  phosphors;  i.e.,  perceptible  tenebrescence  requires  about  100  times  as  much 
cathode-ray  excitation  energy  as  is  required  to  produce  perceptible  luminescence. 

18.  SPECIFIC  CHARACTERISTICS  OF  USEFUL  PHOSPHORS 
AND  SCOTOPHORS 

Phosphors  and  scotophors  may  be  considered  as  high-frequency  transformers  which  can 
transform  a  megavolt-wide  range  of  invisible  photon  or  particle  energies  into  emission  or 
absorption  of  light  in  or  near  the  1.5-volt-wide  visible  region  of  the  electromagnetic  spec- 
trum. The  output  frequencies  of  these  materials  are  in  the  range  from  about  4  X  1014 
to  S  X  1014  cycles  per  second  (red  to  violet). 

The  chief  commercial  uses  of  phosphors  are  in  electron  discharge  devices,  such  as  fluores- 
cent lamps  and  cathode-ray  tubes  for  radar  or  television.  Fluorescent  lamps  contain 
mercury  vaapor  at  a  pressure  of  about  4  M,  which,  under  electron  excitation,  emits  invis- 
ible 2537  A  radiation  which,  in  turn,  excites  visible  luminescence  in  an  internal  phosphor 
coating.  White-emitting  coatings  for  fluorescent  lamps  are  made  by  mechanically  mixing 
blue-emitting  magnesium  tungstate  and  yellow-emitting  manganese-activated  zinc- 
beryl  Hum-silicate  phosphors  which  afford  quantum  efficiencies  of  the  order  of  90  per  cent 
and  overall  efficiencies  four  times  as  great  as  most  incandescent  lamps. 

Cathode-ray-tube  screens  comprise  a  much  greater  range  and  variety  of  materials  than 
those  used  in  fluorescent  lamps.  Table  2  shows  some  approximofe  general  properties  of  the 
more  useful  cathode-ray-tube  screens,  including  all  those  presently  coded  by  the  Radio  Man- 
ufacturers Association  (RM  A) .  The  screens  are  arranged  in  the  order  of  their  approximate 
persistences  after  excitation  by  6-kv  cathode  rays  at  about  1  microampere  per  square 
centimeter.  The  dark-trace  screen  at  the  right  of  Table  2  exhibits  decelerated  decay  as 
the  excitation  density  is  increased;  the  sulfide-type  bright-trace  phosphor  screens  exhibit 
accelerated  decays  with  increasing  degree  of  excitation.  By  observing  a  tenebrescent 
image  at  some  time  after  excitation,  it  is  possible  to  obtain  greater  relative  contrast  between 
image  traces  than  during  the  excitation  interval.  Conversely,  the  normal  bright-trace 
screens  may  be  used  as  contrast  or  ratio  compressors. 


15-38 


ELECTRO-OPTICAL   DEVICES 


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SPECIFIC  CHARACTERISTICS  OF  USEFUL  PHOSPHORS      15-39 


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15-40 


ELECTRO-OPTICAL  DEVICES 


Wavelength  vs  energy  plot? 
32  100*32        45 


Representative  spectral  distribution  curses  of  the  major  cathode-ray-tube  screens  are 
shown  in  Fig.  10.  The  numbers  near  the  peaks  of  the  emission  bands  of  individual  phos- 
phors are  the  heights  of  the  bands  relative  to  the  peak  of  the  cathodoluminescence  band 
of  a-Zn2SiO4:Mn  (PI)  which  is  arbitrarily  set  equal  to  100.  The  visible  cathodolumines- 
cence efficiency  of  conventional  Pi  screens  is  about  3  candles  per  watt  at  6  kv  and  1  micro- 
ampere per  square  centimeter.  The  spectral  distribution  curves  of  the  white-emitting 
P4  (monochrome-television)  and  P6  (color-television)  screens  may  vary  considerably 
among  different  manufacturers,  since  there  are  many  possible  ways  of  producing  equiva- 
lent-appearing white  colors.  The  optimum  white  is  yet  to  be  decided  by  popular  approval, 
although  P4  screens  are  presently  standardized  at  a  color  temperature  of  about  6500-7000 
deg  Kelvin.  The  P4  screen  is  usually  a  two-component  mixture  of  blue-emitting  and 
yellow-emitting  phosphors  [all-sulfide(selenide)  or  sulnde(selenide) -silicate];  the  P6 

screen  is  usually  a  three-component 
mixture  of  blue-,  green-,  and  red-emit- 
ting sulfide-  (selenide)  phosphors. 

The  optimum  screen  thicknesses  for 
cathode-ray  tubes  vary  from  about  1 
mg/cm2  for  phosphors  having  average 
particle  sizes  less  than  1  micron,  to  22 
mg/cm2  for  cascade-screen  (P7  and 
P14)  phosphors  having  average  parti- 
cle sizes  near  15  microns.  The  evapo- 
rated P10  (KC1)  screen  thickness  is 
about  12  microns.  These  thicknesses 
are  for  operation  below  about  10  kv 
and  must  be  increased  for  higher  volt- 
ages according  to  eqs.  (10)  and  (11). 
The  strontium-containing  phosphors 
listed  at  the  bottom  of  Table  1  are 
infrared-stimulable  materials  which 
have  quite  deep  electron  traps  (of  the 
order  of  1  electron  volt) .  These  mate- 
rials can  store  most  of  their  latent 
phosphorescence  energy  for  several 
days,  even  at  room  temperature.  Effi- 
cient release  of  the  stored  phosphores- 
cence energy  is  obtained  by  irradiating 
the  previously  excited  material  with 
near-infrared  which  may  be  varied  in 
intensity  to  alter  the  normal  concave- 
upward  decay  of  the  phosphor.  These 
phosphors  are  good  photolumineseent 
materials  but  are  relatively  inefficient 
under  cathode  rays.  By  cocrystalliz- 
ing  sulfate  ions  in  copper-activated 
zinc  sulfide  phosphor,  an  unusual 


FIG.  10.    Representative  Spectra  of  Seine  Cathode-ray- 
tube  Screens.     P4  and  P6  screens  vary  considerably 
among  different  manufacturers. 


infrared-quenchable  phosphor  is  produced  which  affords  long  phosphorescence  plus  the 
ability  to  decrease  or  terminate  the  phosphorescence  by  quenching  with  infrared  at  an 
arbitrary  instant  during  the  decay  time.  Infrared-sensitive  phosphors  may  be  used  to 
store  information  for  controllable  time  intervals  or  to  convert  positive  images  into  negative 
images  (or  vice  versa) . 

The  intensity  and  duration  of  phosphorescence  after  cathode-ray  excitation  is  generally 
less  than  that  obtained  after  excitation  by  photons.  Cathode-ray-excited  phosphorescence 
may  be  increased  by  using  the  cascade  principle.  The  cascade  principle  involves  construct- 
ing a  stratified-layer  screen  wherein  an  efficient  photophosphorescent  material  is  covered 
with  a  layer  of  an  efficient  cathodoluminescent  material  whose  emission  band  overlaps  the 
excitation  band  of  the  photophosphorescent  material.  By  this  method,  cathode-ray 
energy  is  intermediately  converted  into  photon  energy  which  is  more  efficient  in  exciting 
phosphors  to  produce  phosphorescence.  Cascading  may  be  used  also  to  excite  cathode-ray- 
modulated  photoluminescence  in  materials,  such  as  the  infrared-stimulable  phosphors, 
which  are  inefficient  under  direct  cathode-ray  excitation.  The  P7  and  P14  screens  are 
examples  of  practical  cascade  screens  which  were  devised  for  radar  cathode-ray  tubes. 


ELECTRON  GUN 


15-41 


BIBLIOGRAPHY 

1.  Leverenz,  H.  W.,  Cathodoluminescence  as  Applied  in  Television,  R,€.A.  Rev.,  VoL  V,  No.  2,  131- 

176  (October  1940). 

2.  Leverenz,  H.  W.,  Luminescence  and  Tenebrescence  as  Applied  in  Radar,  R.C.A.  Rei.,  VoL  VII, 

No.  2,  199-240  (June  1946). 

3.  Some  of  the  data  mentioned  herein  were  obtained  during  the  coarse  of  Contracts  NDCrc-150 

and  OEMsr-440  between  the  Office  of  Scientific  Research  and  Development  and  the  Radio 
Corporation  of  America.  A  more  complete  account,  •with  over  400  further  references,  may  be 
found  in  H.  W.  Leverenz,  Final  Report  on  Research  and  Development  Leading  to  New  and 
Improved  Radar  Indicators,  Report  NDRC-1 4-498,  June  30,  1945,  obtainable  from  the  Office 
of  the  Publication  Board,  Department  of  Commerce,  Washington,  D.  C.  (PB254S1). 

4.  Leverenz,  H.  W.,  Luminescent  Solids  (Phosphors),  Science,  VoL  109,  No.  2826,  183-195  (Feb.  25, 

1949). 

5.  Leverenz,  H.  W.,  An.  Introduction  to  Luminescence  of  Solids,  Wiley  (1950). 

6.  Fonda,  G.  R.,  and  Seitz,  F.  (Editors),  Preparation  and  Characteristic*  of  Solid  Lumw&cent  Male- 

rials,  Wiley  (1948). 


CATHODE-RAY  TUBES 

By  L.  E.  Swedlund 


The  cathode-ray  tubes  considered  herein  are  those  electron  tubes  in  which  a  relatively 

low-current  electron  beam  of  small  circular  cross-section  is  focused  on,  and  deflected 
across,  a  luminescent  screen.    Their 

Accelerating       First          Second  _- v 

electrode        artode         anode  ^^^^^^^        ^ 

x    \    ~f  Luminescent 
JJJ            « .  — 

IjuM   


Grid 


Gross  over 


Masking 
aperture 


T 


principal  applications  are  in  tele- 
vision receivers,  oscillographs,  and 
xadar-type  indicators.  Their  great 
value  is  the  ability  to  display  varia- 
tions in  voltage  and  current  without 
the  limitations  of  mechanical  inertia, 
and  their  relatively  low  cost.  Except 
for  a  few  specialized  designs,  they  are 
sealed-off,  high-vacuum  tubes,  hav- 
ing an  electron,  gun  with  a  heater- 
type,  oxide-coated  cathode  located 
in.  the  neck  of  a  cone-shaped  glass 
bulb.  A  luminescent  screen,  is  deposited  in  the  large,  nearly  flat  end  of  this  cone.  Figure  1 
shows  an  electrostatic-focus,  electrostatic-deflection  oscillograph-type  cathode-ray  tube, 
and  Fig.  2  a  magnetic-focus,  magnetic-deflection  tube,  for  television  image  reproduction. 
Magnetic  focus  is  very  seldom  used  with  electrostatic  deflection,  but  electrostatic  focus 
is  often  used  with  magnetic  deflection.  The  principal  parts  of  the  cathode-ray  tube  are 
the  electron  gun,  the  bulb,  and  the  luminescent  screen. 


FIG.  1.    Hectrostatic-focus  Electrostatic-deflection  Cath- 
ode-ray Tube 


19.  ELECTRON  GUN 


The  design  of  the  electron  gun  is  based  on  the  principles  of  electron  optics  outlined  in 
Section  14.    However,  owing  to  the  complex  nature  of  the  electron  paths,  particularly  in 

the  region  of  the  grid  and  cathode,  and  to  the 
large  number  of  interrelated  factors,  much  of 
the  design  is  based  on  experimental  informa- 
tion. The  active  cathode  surface  is  a  small, 
flat,  oxide-coated  nickel  surface  normal  to  the 
gun  axis.  Directly  heated,  high-temperature 
cathodes  are  occasionally  used  in  high-voltage 
tubes  to  withstand  better  the  effects  of  positive- 
ion  bombardment.  A  control  grid  having  a 
round  aperture  of  approximately  <XG4-ih.  diam- 
eter is  spaced  as  closely  as  practical  to  the 
cathode.  As  the  spacing  may  amount  to  only 
0.002  in.  when  the  cathode  is  hot,  it  is  difficult 
to  hold  this  spacing  constant  from  tube  to  tube.  This  small  variation  in  spacing  results 
in  a  fairly  large  variation  in  control-grid  voltage  for  beam-current  cutoff.  The  control  grid 
is  nearly  always  designed  to  operate  at  a  negative  potential  with  respect  to  the  cathode. 
The  beam  is  accelerated  by  means  of  a  screen  grid,  first  anode,  and  second  anode;  by  a 
screen  grid  and  anode;  or  directly  by  an  anode  to  full-beam  potential.  In  each  case,  the 


Cath&de  / 
Cross-over 


FIG.    2.     Magnetic-focus    Magnetic-deflection. 
Cathode-ray  Tube 


15-42  ELECTRO-OPTICAL  DEVICES 

spacing  between  the  first  accelerating  electrode  and  the  control  grid  is  adjusted  to  provide 
about  the  same  accelerating  field  at  the  cathode.  Maximum  beam  currents  of  the  order 
of  1  milliampere  are  drawn  at  zero  bias,  and  the  specified  control-grid  bias  for  beam  eutofT 
varies  from  approximately  20  to  100  volts. 

OPERATION  OF  THE  ELECTRON  GUN.  The  field  above  the  cathode  draws  the 
electrons  into  a  focus  in  a  few  millimeters'  travel.  In  this  field  they  cross  over,  and  beyond 
it  they  travel  in  straight  paths  until  they  enter  the  final  focusing  field  near  the  end  of  the 
gun.  Here  the  electrons  are  made  to  converge  to  a  second  focus  or  cross-over  at  the 
screen.  The  final  focusing  field  or  electron  lens  may  be  either  electrostatic  or  magnetic. 
The  electrostatic  lens  consists  of  coaxial  cylinders,  apertured  disks,  or  a  combination  of 
both.  These  components  are  always  mounted  inside  the  tube.  The  magnetic  lens  is 
always  mounted  outside  the  neck  of  the  tube  and  usually  is  a  coaxial  coil  encased  in  an 
iron  shell  except  for  a  short  axial  magnetic  gap  on  the  inside  surface.  In  both  cases  a 
very  good  degree  of  rotational  symmetry  is  required.  The  usual  error  due  to  any  lack  of 
symmetry  is  astigmatism,  which  results  in  an  elliptical  shape  when  the  spot  is  focused. 
Electrostatic  focus  is  adjusted  by  varying  the  first-anode  voltage.  The  first-anode  voltage 
is  usualb'  about  one-fifth  the  second-anode  voltage.  Magnetic  focus  is  adjusted  by  chang- 
ing the  magnetic  field  in  the  region  of  the  electron  beam.  This  is  done  by  varying  the 
current  in  an  electromagnetic  focus  coil  or  by  varying  a  magnetic  shunt  in  a  permanent- 
magnet  focus  unit. 

ELECTROSTATIC  DEFLECTION.  Figure  1  shows  two  pairs  of  deflecting  plates  at 
right  angles  to  each  other  at  the  exit  of  the  electron  gun.  The  average  potential  of  each 
pair  should  be  near  that  of  the  second  anode  of  the  gun  since  a  difference  will  produce  a 
focusing  effect.  Since  such  focusing  produces  a  strong  astigmatic  effect  it  is  sometimes 
possible  to  adjust  their  average  potential  to  counteract  astigmatism  in  the  electron  gun. 
Most  tubes,  however,  are  so  well  made  that  such  correction  is  not  needed.  The  best  uni- 
formity of  focus  is  attained  with  symmetrical  or  push-pull  deflection.  It  is  possible  to 
connect  one  plate  of  a  pair  to  the  second  anode  and  apply  voltage  only  to  the  other  at 
the  expense  of  focus  and  sometimes  linearity  of  deflection.  When  the  deflection  plates 
are  parallel  the  deflection  can  be  calculated  from  the  following  equation. 

-38 

where  h  is  the  deflection  from  the  center,  L  is  the  plate  to  screen  distance,  I  the  deflecting- 
plate  length,  Vd  the  difference  in  potential  between  the  plates,  Vb  the  second-anode  or 
beam  voltage,  and  d  the  spacing  between  plates.  In  order  to  provide  maximum  sensi- 
tivity, the  plates  are  usually  shaped  to  follow  the  contour  of  the  deflected  beam.  The 
expression  for  deflection  becomes  more  complex  but  has  about  the  same  proportionality 
factors. 

MAGNETIC  DEFLECTION.  Magnetic  deflection  is  accomplished  by  placing  a  mag- 
netic field  at  right  angles  to  the  beam  path.  This  field  is  usually  provided  by  a  pair  of 
coaxial  coils  placed  on  opposite  sides  of  the  neck  in  order  to  produce  a  nearly  uniform 
deflecting  field.  The  beam  moves  at  right  angles  to  the  magnetic  field  in  a  direction 
indicated  by  the  familiar  left-hand  motor  rule,  remembering,  however,  that  conventional 
current  flow  is  opposite  that  of  electrons.  Unlike  electrostatic  deflection,  it  is  possible  and 
advantageous  to  place  a  second  pair  of  coils  for  orthogonal  deflection  in  the  same  region 
as  the  first.  A  combination  of  deflection  coils  is  known  as  a  deflection  yoke  and  is  ordi- 
narily designed  to  meet  the  rather  specialized  requirements  of  television.  Magnetic  pole 
pieces  and  return  paths  are  sometimes  provided  to  decrease  the  current  required  by  reduc- 
ing the  reluctance  of  the  magnetic  path.  The  deflection  h  at  the  screen  for  a  magnetic 
field  of  length  Z  having  uniform  flux  density  of  B  at  a  distance  L  from  the  screen,  and 
for  a  final  voltage  J?&,  is 

,        0.298  X  W^BIL 

h  •• — (2) 

VVb 

Since  there  is  a  large  air  gap  in  these  coils,  the  flux  density  is  proportional  to  the  current 
and  thus  deflection  is  proportional  to  the  current.  Although  some  approximations  have 
been  made,  this  equation  holds  well  up  to  60°  total  deflecting  angle.  It  is  seldom  desired 
that  the  field  be  uniform  across  the  neck  because  the  screen  is  usually  not  properly  curved 
to  produce  a  pattern  which  looks  rectangular.  In  the  reflective  optics  projection  system, 
for  example,  the  pattern  is  made  to  "barrel"  slightly  to  correct  for  "pin-cushion"  distortion 
in  the  optics  (see  Section  14) . 

COMPARISON  OF  MAGNETIC  AND  ELECTROSTATIC  DEFLECTION.  In  com- 
paring eqs-  (1)  and  (2)  for  electrostatic  and  magnetic  deflection,  it  is  to  be  noted  that, 
with  an  increase  of  beam  voltage,  deflection  decreases  with  the  anode  voltage  in  the  former 


BULBS  FOR  CATHODE-RAY  TUBES  15-43 

and  as  the  square  root  of  the  anode  voltage  in  the  latter.  Thus,  magnetic  deflection  be- 
comes relatively  more  favorable  as  the  voltage  is  raised.  In  television,  this,  plus  relatively 
better  focus  and  economic  factors,  sets  the  dividing  point  at  about  5  kv.  Electrostatic  and 
magnetic  deflection  are  occasionally  combined  in  one  tube,  though  tubes  designed  spe- 
cifically for  this  type  of  operation  have  not  found  many  applications. 

DEFLECTION  DEFOCUSING.  Deflection  is  accompanied  by  a  deterioration  of  focus 
known  as  deflection  defocusing.  It  appears  as  an  elongation  of  the  spot  in  the  direction  of 
deflection  in  electrostatic  deflection  and  as  an  elongated  and  rotated  spot  in  magnetic 
deflection.  It  increases  approximately  as  the  square  of  the  deflecting  angle.  Thus  deflec- 
tion distortion  limits  the  deflecting  angle  and  indirectly  the  beam  current  because  the 
amount  of  current  which  can  be  focused  into  a  given  size  spot  is  determined  by  the  size 
of  the  beam  as  it  leaves  the  gun.  For  high  currents  and  wide  deflecting  angles  the  deflec- 
tion distortion  is  five  to  ten  times  as  great  with  electrostatic  deflection  as  with  magnetic 
deflection.  As  a  result,  electrostatic-deflection  tubes  are  designed  for  smaller  focused 
currents  and  narrower  deflecting  angles  than  magnetic-deflection  tubes.  The  change  of 
focus  due  to  the  increase  of  gun  to  screen  distance  with  deflection  is  small  compared  to 
deflection  defocusing  even  with  fiat  screens.  But,  since  deflection  defocusing  also  produces 
a  shortening  of  the  image  focal  distance,  it  is  possible  to  make  a  small  amount  of  compensa- 
tion by  slightly  underf ocusing  the  beam  at  the  center  of  the  screen.  This  effect  is  usually 
so  small  that  it  is  not  worth  while  to  modulate  the  focus  voltage  in  synchronism  with  scan- 
ning to  improve  the  uniformity  of  focus, 

20.  BULBS  FOR  CATHODE-RAY  TUBES 

The  bulb  of  the  cathode-ray  tube  must  meet  specialized  requirements  for  mechanical 
strength,  dimensions,  optical  quality,  and  electrical  insulation.  Special  grades  of  glass  are 
needed  to  meet  these  demands.  Although  the  bulb  has  to  withstand  only  atmospheric 
pressure  (15  Ib  per  sq  in.)  it  is  good  practice  to  design  for  a  breaking  strength  of  60  Ib  or 
more  per  sq  in.  This  is  done  to  avoid  failure  due  to  slight  scratches,  temperature  shock, 
and  aging.  Great  care  should  be  exercised,  particularly  in  handling  the  larger  tubes,  to 
protect  personnel  from  flying  glass  in  the  event  of  accidental  implosion.  Generally  the 
zone  of  highest  stress  in  a  bulb  is  near  the  rim  of  the  face.  Here  the  outside  surface  of  the 
glass  is  in  tension,  in  which  it  is  weak,  and  so  particular  care  should  be  taken  to  avoid 
scratches  and  shocks  in  this  zone.  In  the  past  large  bulbs  were  usually  made  of  heat- 
resisting,  low-thermal-expansion  glasses,  but  because  of  both  cost  and  optical  quality  most 
bulbs  are  now  made  of  a  soft  glass.  For  all  but  the  low-voltage  oscillograph-type  tubes, 
this  is  a  lead-type  glass  in  contrast  to  the  usual  lime-type  soft  glass.  It  is  chosen  to  obtain 
high  electrical  resistivity.  Large  bulbs  are  mass-produced  by  pressing  parts  which  are 
fused  together.  If  the  surface  must  be  very  good  and  accurately  shaped,  as  in  projection 
type  tubes,  the  face  plate  may  be  ground  and  polished  before  sealing  to  the  bulb. 

LIMITATIONS  IMPOSED  BY  THE  LUMINESCENT  SCREEN.  The  luminescent 
screen  is  a  very  important  part  of  the  cathode-ray  tube.  (See  articles  15—18.)  The  limita- 
tions and  requirements  of  the  luminescent  screen  have  an  important  bearing  on  gun-design 
problems.  Both  the  amount  of  current  that  can  be  sharply  focused  and  the  amount  that 
can  be  utilized  efficiently  by  the  screen  are  limited  at  low  voltage.  High  brightness, 
consequently,  has  to  be  obtained  by  raising  the  beam  voltage  applied  to  the  screen.  Al- 
though raising  the  anode  voltage  improves  the  focus  and  screen  efficiency,  it  also  increases 
electrical  insulation  problems,  introduces  screen  charging,  and  requires  higher  deflection 
energy.  Specially  developed,  low-current,  high-voltage  power  supplies  which  have  low 
cost,  compactness,  and  safety  have  made  this  problem  less  difficult. 

SCREEN  SIZE  AND  BRIGHTNESS.  When  the  screen  size  is  increased  without  chang- 
ing the  gun  or  deflecting  angle,  the  anode  voltage  must  be  raised  to  maintain  the  same 
number  of  lines  resolution  and  brightness.  This  is  due  to  the  fact  that  the  size  of  the 
focused  spot  increases  with  bulb  length  and  that  the  beam  energy  per  unit  of  screen  area 
determines  the  brightness.  Projection  tubes  require  the  highest  screen  brightness  and  are 
therefore  operated  at  the  highest  voltages.  Luminescent  screens  are  very  thin  and  thus 
have  a  very  small  thermal  storage  capacity.  Great  care  must  be  taken  not  to  move  the 
beam  slowly  or  over  too  small  an  area.  With  a  small  overload  the  screen  may  be  damaged 
only  temporarily,  as  by  overheating,  or  to  a  light  "burn"  which  gradually  fades  out.  The 
total  time  of  screen  bombardment  during  the  life  of  a  tube  is  surprisingly  small.  Consider  a 
television  tube  scanned  with  a  525-line  raster.  This  provides  roughly  360,000  picture 
elements;  and  so,  neglecting  blanked-out  return  line  time  and  modulation,  each  element 
of  the  picture  area  is  being  bombarded  only  l/360,000th  of  the  time.  One  thousand  hours 
amounts  to  3,600.000  seconds.  Thus  in  a  representative  tube  lifetime,  the  screen  has  been 


15-44  ELECTRO-OPTICAL  DEVICES 

bombarded  less  than  10  seconds.  However,  during  this  short  time  it  is  bombarded  at  a 
very  high  momentary  energy  density;  for  example,  in  a  5-in.  projection  tube  it  amounts 
to  25  kw/cm2. 

DISCHARGE  OF  SCREEN.  Luminescent  screen  materials  are  good  electrical  insulators 
and  are  prevented  from  charging  up  negatively  under  electron  bombardment  only  by 
virtue  of  a  secondary-to-primary  electron-emission  ratio  equal  to  or  greater  than  unity. 
The  range  of  voltage  where  this  is  true  varies  with  the  type  of  screen  material,  its  manu- 
facture, and  application.  Under  favorable  conditions  screens  can  be  discharged  by  second- 
ary emission  over  the  500  to  25,000  volt  range.  At  the  upper  end  of  the  range  not  all 
phosphors  can  be  used  and  a  charge  of  several  thousand  volts  may  build  up  on  the  screen, 
thus  reducing  the  effective  screen  voltage.  In  general,  secondary  emission  becomes  poorer 
at  low  current  density  and  with  use.  Poor  secondary  emission  at  low  voltage  causes  the 
screen  to  charge  up  enough  to  deflect  the  beam  off  the  screen.  For  example,  trouble  is 
likely  to  occur  if  beam  current  flows  to  the  screen  while  the  anode  voltage  is  building  up. 
A  charge  is  then  built  up  before  the  screen  is  bombarded  with  full-voltage  electrons.  At 
the  upper  end  of  the  range  the  light  output  fails  to  increase  at  the  normal  rate  with  voltage 
increase  and  the  pattern  may  become  unstable.  The  effects  of  poor  secondary  emission 
may  be  overcome  by  mounting  the  phosphor  on  a  conducting  surface  and,  above  about 
7000  volts,  by  placing  a  very  thin,  light  metal  back  on  the  luminescent  screen.  The  metal 
back  is  especially  desk-able  because  it  can  be  made  light-reflecting  in  order  to  utilize  the 
light  ordinarily  lost  through  the  back  of  the  screen.  It  can  also  serve  as  an  effective  barrier 
to  negative  ions. 

ION  SPOT  IN  SCREEN.    Ion  spot  in  the  screen  is  a  problem,  particularly  in  magnet- 
ically deflected  tubes.    It  is  the  area  in  the  luminescent  screen  which  has  been  damaged 
by  negative-ion  bombardment.    The  screen  is  damaged  almost  altogether  by  the  negative 
ions  which  form  in  or  near  the  cathode  surface  when  it  is  bombarded  by  positive  ions.   The 
.  rate  of  damage  increases  with  voltage,  but  it  is 

* most  noticeable  at  about  7  kv  because  above  this 

voltage  the  electrons  begin  to  penetrate  below 
the  damaged  surface  layer  of  the  screen.  It  is 

^t^Tilft     Electrons?          ' greatly  affected  by  the  amount  and  kind  of  gas 
.J-fffF" j         j  22L5!Jj!e     left  in  the  tube,  but  it  is  almost  impossible  to 

eliminate  ion  damage  completely  by  evacuation 
and  gas  getters.  Because  electrostatic  focusing 
is  independent  of  mass  and  magnetic  focusing  is 
not,  electrostatic  focus  guns  focus  the  ions,  thus 
forming  small  ion  spots  rather  quickly  whereas 
magnetic  focus  tubes  form  large  ion  spots  slowly. 
Initially  the  small  spot  is  more  objectionable,  but 
with  time  the  large  spot  becomes  more  objection- 
able. In  addition  to  the  metal-backed  screen, 
which  is  rather  expensive  to  apply  to  low-voltage 

B.  "Tilted  lens"  gun  *  tubes,  there  is  another  effective  solution  known 

FIG.  3.    Sectional  Views  of  Ion  Trap  Guns    *»  **  "io11  W  electron  gun.    Figure  3  shows 

two  constructions  of  this  idea.  Both  are  simpli- 
fied modifications  of  the  mass  spectrograph.  The  combined  beam  of  ions  and  electrons  is 
directed  off  the  center  of  the  gun  into  a  trap  in  A  by  bending  the  anode  and  accelerator 
and  in  B  by  means  of  a  tilted  electrostatic  lens.  In  both,  a  magnetic  field  is  applied  to 
restore  the  electrons  to  the  axis.  The  secondary  magnetic  field  used  in  construction  B  is 
useful  not  only  in  aligning  the  beam  with  the  gun  axis  but  also  in  overcoming  the  slight 
mechanical  misalignments  that  may  be  present. 

21.  CHARACTERISTICS  OF  THE  IMAGE 

SPOT  SIZE.  The  image  detail  or  resolution  is  determined  principally  by  the  size  of 
the  focused  electron  image  on  the  screen.  Contrast  and  gram,  size  of  the  screen  may  also 
be  factors.  The  spot  does  not  have  a  definite  edge  but  decreases  in  brightness  from  the 
center  approximately  as  a  segment  of  a  sine  wave.  When  the  beam  scans  a  television-type 
raster  it  is  found  that,  when  the  lines  overlap  to  the  point  where  the  brightness  is  half  that 
of  the  center,  the  line  structure  practically  disappears,  For  this  reason  it  is  convenient  to 
express  spot  size  as  the  width  at  these  half-brightness  values.  This  is  the  basis  of  the 
"compressed  raster"  method  of  measuring  spot  size  sometimes  used  for  oscillograph-type 
fetfees.  However,  the  rate  at  which  the  brightness  falls  off  toward  the  edge  of  the  spot 
is «pGrfcaa?fc  m  determining  the  resolving  power,  and  so  for  television  reproduction  tubes  a 


CHARACTERISTICS  OF  THE  IMAGE  15-45 

resolution  pattern  signal  which  takes  this  into  account  is  generally  used.     (See  Fig.  10, 
p.  15-25.) 

The  resolution  pattern  has  sets  of  spaced  black  and  white  lines  usually  converging  in 
wedges  along  with  circles  and  bars  for  other  tests.  The  limiting  resolution  is  denned  as 
the  point  where  the  black  and  white  lines  merge.  It  is  important  that  the  black  to  white 
grid  signal  level  be  maintained  beyond  the  value  of  resolution  being  read.  If  the  reproduc- 
tion tube  is  not  to  limit  the  resolution  of  the  television  signal  noticeably,  its  limiting 
resolution  should  be  about  three  times  that  of  the  television  signal.  The  resolution  is  best 
at  the  center  of  the  raster  and  poorest  in  the  corners,  and  since  it  depends  on  the  deflection 
yoke  it  may  vary  with  the  direction  of  scanning.  Its  value  is  usually  expressed  in  terms 
of  the  number  of  black  and  white  lines  required  to  fill  the  vertical  height  of  the  raster. 
In  photographic  and  optical  literature  it  is  generally  the  practice  to  count  only  the  number 
of  black  lines  per  unit  of  length. 

CONTRAST.  The  ability  to  see  an  image  on  a  screen  depends  not  only  on  the  bright- 
ness of  the  trace  but  also  on  the  ratio  of  this  brightness  to  the  background  brightness.  This 
is  known  as  the  brightness  contrast  ratio.  This  ratio,  also  sometimes  referred  to  as  con- 
trast range,  determines  the  number  of  distinct  half  tones  which  can  be  reproduced.  Its 
value  depends  not  only  on  the  inherent  characteristic  of  the  screen  but  also  upon  the  area 
and  distribution  of  illumination  in  the  screen.  The  minimum  or  limiting  value  of  contrast 
ratio  can  be  found  by  uniformly  scanning  all  but  a  small  black  spot  in  the  center  of  the 
raster.  A  contrast  ratio  representative  of  large  areas  can  be  found  by  scanning  half  the 
picture  area  and  measuring  the  brightness  in  the  center  of  each  half. 

GAMMA.  The  term  gamma  is  borrowed  from  photographic  practice  to  designate  the 
half-tone  characteristics  of  a  television  system.  Over  the  brightness  range  useful  for  tele- 
vision reproduction  the  eye  recognizes  brightness  differences  as  percentage  increases  in 
steps  of  1  to  2  per  cent.  For  this  reason  it  is  convenient  to  plot  the  grid  drive  versus  light 
output  of  a  picture  tube  on  log-log  scales.  The  slope  of  this  curve  is  known  as  the  gamma. 
In  the  same  way  each  part  of  the  system  from  the  light  in  the  scene  on  through  the  ampli- 
fiers has  a  gamma  value.  The  overall  gamma  is  the  product  of  those  of  the  series  of  ele- 
ments. Unity  overall  gamma,  which  is  usually  desired,  then  represents  brightness  values 
in  the  image  proportional  to  those  in  the  scene.  The  gamma  should  be  uniform  over  the 
whole  brightness  range  for  accurate  reproduction,  but  it  is  usually  less  at  the  low  and  high 
brightness  levels,  as  is  also  true  in  photographic  reproduction.  It  is  the  practice  in  photog- 
raphy to  refer  to  gamma  as  "contrast."  This  should  not  be  confused  with  contrast  ratio. 

It  has  become  the  practice  to  mark  the  d-c  bias  and  the  signal  controls  of  television 
receivers  as  "brightness"  and  "contrast"  respectively.  However,  both  controls  affect 
the  gamma  and  brightness.  Overbiasing  increases  gamma  by  eliminating  detail  in  the 
shadows;  underbiasing  decreases  it  by  "washing  out"  shadow  detail.  Average  brightness 
varies  with  d-c  bias.  Increase  in  brightness  by  signal  increase  is  usually  limited  by  the 
increase  in  spot  size  or  amplifier  overload.  This  loss  of  detail  and  contrast  in  the  highlights 
is  sometimes  referred  to  as  "blooming."  In  a  picture  tube  the  average  gamma  varies  from 
3  to  2,  depending  on  the  amount  of  masking  in  the  gun.  Correction  to  match  the  gamma 
of  a  standard  transmitted  signal  must  be  done  by  introducing  suitable  curvature  in  the 
amplifier  input  versus  output  characteristic. 

POST  ACCELERATION.  The  beam  is  sometimes  accelerated  after  deflection  in  order 
to  increase  the  spot  brightness  without  a  proportionate  increase  in  deflection  energy.  For 
this  purpose  the  bulb  conductive  coating  is  separated  and  brought  out  to  a  third  anode 
terminal.  It  has  been  applied  only  to  electrostatic  deflection  tubes  because  little  or  no 
gain  results  when  it  is  used  with  magnetic  deflection.  Post  acceleration  results  in  a  focus- 
ing action  which  tends  to  bend  the  deflected  beam  back  to  the  axis.  There  is  also 
a  small  amount  of  focusing  action  on  the  spot  which  is  readily  corrected  by  raising 
the  first  anode  voltage  a  qm»-fl  amount.  In  a  typical  post-accelerator-type  oscillograph 
tube  the  deflection  voltage  must  be  raised  20  per  cent  when  the  third  anode  is  in- 
creased from  equal  to  double  the  second  anode  voltage.  This  increase  in  deflection  voltage 
is  accompanied  by  an  increase  in  deflection  defocusing  so  that  the  sharpness  of  focus  is 
not  as  good  as  an  equivalent  non-accelerator  tube  whose  screen  voltage  is  the  same.  The 
deflection  sensitivity  also  will  not  be  as  much  less  as  indicated  on  the  comparative  tests 
on  the  accelerator  tube  because  the  deflection  plates  can  be  designed  for  closer  spacing. 
However,  the  gain  in  brightness  and  deflection  sensitivity  along  with  the  fact  that  the  high- 
voltage  power  supply  is  grounded  between  its  negative  and  positive  terminals  often  justifies 
the  higher  cost  of  a  tube  of  this  type.  The  accelerator  ratio  is  ordinarily  limited  to  about 
2,  but  where  very  high  spot  brightness  is  needed  tubes  may  be  designed  to  operate  up  to 
a  ratio  of  10  times. 

PHOTOGRAPHY  OF  CATHODE-RAY-TUBE  TRACES.  A  permanent  record  is 
sometimes  needed,  and  when  a  tracing  of  the  pattern  on  the  tube  cannot  be  readily  made 


15-46 


ELECTRO-OPTICAL  DEVICES 


it  can  usually  be  photographed.  Unless  a  time  exposure  can  be  made  the  brightness  level 
suitable  for  viewing  must  usually  be  increased  by  raising  the  beam  current  and  possibly 
by  raising  the  screen  voltage.  Luminescent  screens  such  as  the  P5  and  Pll  are  particularly 
suited  to  photography.  The  blue  light  of  these  screens  provides  energy  where  the  film  is 
most  sensitive.  The  lower-cost  "color  blind"  films  may  sometimes  serve  nearly  as  well 
as  the  orthochromatic  and  panchromatic  films.  Since  the  Pll  screen  has  approximately 
10  times  the  photographic  and  visual  efficiency  of  the  P5  screen  it  is  preferred,  but  occa- 
sionally its  longer  persistence  may  be  a  limitation.  The  proper  exposure  depends  on  lens 
speed,  film  sensitivity,  screen  size,  spot  brightness,  and  spot  velocity.  Tables  suitable  for 
estimating  the  best  exposure  are  available  from  tube  manufacturers,  but  usually  a  suitable 
value  can  be  found  from  a  few  test  exposures.  Single  television  frames  can  be  photo- 
graphed with  a  hand  camera  having  a  fast  lens  and  by  using  high-speed  panchromatic 
film.  For  motion-picture  records  the  camera  must  be  exposed  in  synchronism  with  the 
frame  frequency. 


22.  TELEVISION  CATHODE-RAY  REPRODUCTION  TUBES 

Cathode-ray  tubes  for  television  image  reproduction  are  known  as  picture  tubes  or 
kinescopes.  In  directly  viewed  tubes  the  size  of  the  screen  largely  determines  the  cost  and 
operating  voltage.  Below  a  diameter  of  7  in.  the  reduction  hi  cost  is  not  great  and  the 
image  size  is  too  small  for  comfortable  viewing.  Ten-,  fifteen-,  and  twenty-inch  bulbs 
provide  images  in  steps  of  double-sized  areas.  Intermediate  sizes  such  as  12  in.  are  also 
used.  In  the  largest  bulbs  the  economic  limit  for  evacuated  glass  bulbs  is  reached.  Larger 
images  have  to  be  produced  by  optical  projection  from  a  small  screen  tube.  This  small 
tube,  known  as  a  projection  tube,  must  produce  a  much  brighter  image  to  illuminate  the 
larger  viewing  screen  and  to  make  up  for  the  losses  in  the  optical  system  and  so  is  operated 
at  a  higher  anode  voltage. 

Table  1.     Television  Picture-reproduction  Cathode-ray  Tubes 


Type 

Size,  in. 

Typical  Operation 

Comments 

Nominal 
Diam- 
eter 

Length 

Heater 

Anode 
2,kv 

Anode 
1,  volts 

Grid 
2, 

volts 

Control 
Grid 
Cutoff 

Focus 

Volts 

Amperes 

5TP4.  .  .  . 

5 

113/4 

6.3 

0.6 

27 

4900 

200 

-70 

ES 

Reflective  optics  pro- 

jection 

7CP4.  .  .  . 

7 

137/lfi 

7 

1365 

250 

-45 

ES 

Monitor  use  in  day- 

light 

7DP4.... 

7 

141/16 

6.3 

0.6 

6 

1430 

250 

-45 

ES 

Small  size  receiver 

7EP4.... 

7 

151/2 

6.3 

0.6 

2.5 

650 

-60 

ES 

ES    deflection,    low- 

cost    receiver;     see 

oscillograph  tubes 

for  deflection  factors 

7JP4.... 

7 

141/2 

6.3 

0.6 

6 

1000 

-120 

ES 

ES  deflection  for  low- 

cost    receiver;     see 

oscillograph  tubes 

for  deflection  factors 

7HP4.... 

7 

13 

6.3 

0.6 

6 

250 

-55 

M 

Small  size  receiver 

IOAP4... 

10 

171/2 

6.3 

0.6 

6 

-60 

M 

Bent  gun  ion  trap 

IOBP4.  .  . 

10 

175/8 

6.3 

0.6 

9 

250 

-45 

M 

Directly  viewed,  tilted 

lens  ion  trap 

10CP4.  .  . 

10 

165/8 

6.3 

0.6 

8 

250 

-45 

M 

Directly  -dewed 

10EP4.  .  . 

10 

175/g 

6.3 

0.6 

8 

250 

-45 

M 

Directly  viewed,  tilted 

lens  ion  trap 

10FP4... 

10 

17 

6.3 

0.6 

9 

250 

-45 

M 

Directly  viewed, 

metal-backed  screen 

I2JP4... 

12 

171/2 

6.3 

0.6 

10 

250 

-45 

M 

Directly  viewed 

15AP4... 

15 

203/16 

6.3 

0.6 

10 

250 

-45 

M 

Large  directly  viewed 

15BP4.  .  . 

15 

19 

6.3 

0.6 

12 

250 

_45 

M 

Large  directly  viewed, 

rectangular  face 

2QBP4.  .  . 

20 

283/4 

6.3 

0.6 

15 

250 

-45 

M 

Large  directly  viewed 

Note  The  above  table  includes  the  tubes  commercially  available.  Developmental  tubes  and  those  recommended  for 
repJacesnent  oaly  are  not  included.  However,  some  of  the  tubes  in  these  lists  are  in  fact  outmoded  and  will  in  time 
beeomse  obsolete.  Owing  to  changes  and  additions  which  are  constantly  being  made,  up-to-date  and  more  detailed 
iilQnaaiatkm  sbauld  be  obtained  from  tube  manufacturers  or  their  agents. 

Tiw*  deflecting  angle  in.  a!i  &e  above  tubes  is  50°  or  slightly  above.    All  use  P4r-type  screens. 


BIBLIOGRAPHY  15-47 

DEFLECTION  AKD  FOCUS  OF  PICTURE  TUBES.  Magnetic  deflection  is  used  for 
all  but  the  smallest  television  picture  tubes  because  it  permits  the  use  of  wider  deflection 
angles  and  higher  beam  currents.  Owing  to  the  desire  for  short  overall  length,  particularly 
in  the  large  tubes,  the  deflecting  angle  is  made  as  large  as  practicable.  Fifty-degree  total 
angle  has  been  found  to  provide  a  good  compromise  between  focus  uniformity,  beam 
current,  and  deflecting  power.  Both  magnetic  and  electrostatic  focus  guns  are  used,  the 
former  being  favored  for  directly  viewed  tubes  and  the  latter  for  project  ion-type  tubes. 
At  the  high  anode  voltages  used  in  the  projection  tubeT  energy  for  the  focusing  coil  becomes 
a  more  important  factor  and  the  coil  is  also  difficuli  to  mount  in  the  reflective  optics  system 
generally  used  for  projection.  The  highest-voltage  power  supplies  tend  to  have  the  poorest 
voltage  regulation,  so  that  a  first  anode  which  can  be  held  in  focus  by  means  of  a  tap  on 
the  bleeder  across  the  high- voltage  supply  is  quite  desirable.  The  bulb  size  of  projection 
tubes  is  limited  by  the  cost  and  bulk  of  the  optical  system  used  with  them.  For  home  use 
a  5-in.  bulb  and  27  kv  appears  to  be  an  economic  choice.  For  intermediate  and  theater- 
size  images  larger  tubes  and  much  higher  operating  voltages  are  justified. 

23.  OSCILLOGRAPH-TYPE  CATHODE-RAY  TUBES 

These  tubes  are  nearly  always  of  the  electrostatic-focus,  electrostatic-deflection  type. 
Their  largest  field  of  application  is  in  low-voltage  oscillographs  for  visual  observations. 
These  operate  in  the  range  of  500  to  2000  volts,  but,  for  the  observation  and  photography 
of  high-frequency  transients,  anode  voltages  up  to  30  kv  may  be  used.  The  higher  bright- 
ness and  sharper  focus  afforded  by  higher-voltage  operation  are  desirable  but  it  results 
in  lower  deflection  sensitivity  and  higher  cost  tubes  and  accessories.  The  signal  must 
usually  be  amplified,  and  because  a  wide  band  width  is  desired  the  gain  per  stage  is  low. 
The  size  of  the  screen  also  influences  the  choice  of  anode  voltage,  since  the  energy  to  the 
screen  should  go  up  with  its  size.  Conversely,  with  a  given  anode  voltage,  a  small-diameter 
tube  may  indicate  more  detail  than  a  large  screen  tube.  Of  the  factors  determining  oscillo- 
graph-tube performance,  deflection  sensitivity  is  the  most  important.  Deflection  sensi- 
tivity is  customarily  expressed  as  the  number  of  millimeters7  deflection  at  the  screen  per 
volt  (d-c)  of  deflection  voltage.  The  reciprocal  of  this  term,  the  deflection  factor,  is  usually 
expressed  in  d-c  volts  per  inch  of  deflection.  The  latter  term  is  favored  for  design  vork. 
Spot  size  and  contrast  are  next  in  importance.  Grid  modulation  sensitivity  is  of  relatively 
less  importance  than  in  television.  It  is  desirable  that  the  focus  remain  sharp  -while  the 
beam  current  is  varied,  so  most  oscillograph-type  tubes  utilize  an  electron  gun  requiring 
essentially  zero-first-anode  current  in  order  to  avoid  the  effects  of  voltage  regulation  in 
the  first  anode  circuit. 

HIGH-FREQUENCY  DEFLECTION.  When  the  deflecting  electrode  leads  must  have 
minimum  inductance,  capacitance,  and  coupling  to  other  leads,  they  are  brought  directly 
through  the  neck  of  the  bulb.  This  is  also  of  value  when  very  wide  frequency  band  ampli- 
fiers are  used.  For  frequencies  of  about  100  megacycles  and  above  it  is  necessary  to  take 
into  account  the  electron  transit  time  through  the  plates.  Special  tubes  with  short-length 
plates  and  high  beam  voltage  have  been  used  to  record  frequencies  as  high  as  10,000  mega- 
cycles per  second. 

MULTIPLE  GUN  OSCILLOGRAPH  TUBES.  Oscfflograph-type  tubes  with  two  or 
three  independent  guns  and  deflection  systems  are  available  for  simultaneously  producing 
two  or  more  traces.  Owing  to  their  more  complex  construction  their  cost  is  higher  than  that 
of  the  equivalent  number  of  single  tubes.  An  electronic  switch  or  the  simultaneous  use  of 
two  or  more  tubes  is  sometimes  satisfactory. 

RADIAL  DEFLECTION  TUBES.  Radial  deflection  tubes  have  a  rod-type  deflecting 
electrode  extending  through  the  face  into  the  middle  of  the  cone.  The  deflection  sensitivity 
of  the  rod  decreases  as  the  square  of  the  deflection  so  that  it  cannot  readily  be  calibrated. 
It  is  used  principally  to  make  a  marker  pip  on  a  circular  time  base. 

BIBLIOGRAPHY 

Law,  R.  R.,  Contrast  in  Kinescopes,  Proc.  I.R.E.,  August  1939,  pp.  311-524. 

Bachman,  C.  H.,  Image  Contrast  in  Television,  Gen.  Elec.  Rev.,  Vol.  48,  No.  9, 13-19  (September  1945). 

Feldt,  Rudolf,  Photographing  Patterns  on  Cathode-ray  Tubes,  Electronics,  February  1944. 


15-48 


ELECTRO-OPTICAL  DEVICES 


Table  2.  Cathode-ray  Tubes  and  Their  Characteristics 

OHciLLoaitApnvnrrH  TUBES 
ELECTROSTATIC-DEFLECTION  TYPES 

Comments 

11     i   8     3?       3 

•a        e!    ,  i-    ss      s 

8        «§     §  i    i*  ^i, 

|           If      &t?      =•§   If  2 

2          c  p     £     u     o       &  £      tS  °tj       £  ^    fl  c  fl  *•• 

i    il  1  1  j  |jj  *|i  ||  mi 

1     ||  1  1  !|  i;    l|l    P  llj| 
1  2  a  a  s-S^  s  s  2!  a  sf  ali-as  g^-S  §sss  g  s  2  g 

Illll!ll1l1ll!lilll-l-5lll-f-gllll 

2'3"3'3*o"5'H"5  £"S"S'S  o  S'S'ti'M^'S'SS  i:3""'S'SiS'"'2's'° 

!3oco3!o3oSo^o5troS  m^  °oos  "oooo 

flowed  by  "A"  indicate  improved  designs  which  are  interchangeable  with  the  old;  however,  the  reverse  ia  not  necessarily  true.  In  the 
on  tubes  it  indicates  the  change  to  zero-first-anode  current  gun. 

MAGNETIODEFLECTION-TYPE  OSCILLOGRAPH  TUBES 

<n 

1 

ndicator 
ndicator 
on  monitor 
ndicator 
ndicator 
ndicator 

Illlll 

a 
P 

oooooo 

s 

.2 
*+3 

a 
O 

'a 
h 

o 

ssasss 

Other 
Screens 
Listed 

ts—  "    ts         ri                    IN     JN" 

Control 
Grid 
Cutoff 

om^tnoow^ 

1  1  I  I  1  1 

Typical  Operation 

idq 

S-S£S¥r3gS?SS§S£S  S3  £  ?SSg  SS5I3 

(N 

•§ 
o 

oooooo 

gpQ 

*  w  ^ 

osk 

j«.^vjcn—  o*n«^«^csooC"~0v(00sON  for>>     ^     tf"30^     «t>—  co 

! 

NO 

fA 

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a°s 

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i  i  1  1  i  i  !  i  1  !  1  i  1  1  M  I      M      1      III      I  1   1  I 

o 

ooo.«o«,o^o*.«o«u^o     o^    «    o«o     o^^r. 

<N 

1 

oooooo 
oooooo 
oooooo 

N 

•§ 

Or>» 
•< 

ooooooooooooooooo     oo    o     ooo     oooo 

OOOOOOOOC300000000      00      0      000      ggOg 

""*"""       ~~           ~      "~"~      ""                                        ~~0 

Heater 

£ 

,0™. 

o 
T3 

Otfi 

<3 

0       0                      00      00      0                                         0 

o     o               oo    oo    o                             o 
o     o                oo     oo     o                              o 

<«•       T                       •«••«•      T^-      >f 

oooooo 

S  E 
1^ 

0>0^^~^0«0^000-^vOOvO      ^>0      xO      0  ONO-0 

00 

'o 

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**>*>*>** 

i-i    ac 

3 

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c 

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0^vOvO<NsOvOsOsO^^O<NOOvOO      OM3      O      sOCNC4      vOvOfS^O 

5 
1 

I 

«S5«  ||  «  3w«^  ^  <  ^^  ^s 

Note:  ItMA  tube  numbers  i( 
case  of  the  electrostatic-deflecti 

Nominal 
Diameter 

.™»J, 

Nominal 
Diameter 

-««««««««««««.«««  w  «  «««  .... 

a 

<jj<              IP; 

_* 

:  :<  '•«*  '•<  '•  '•  •  •  '<<^    '•'•        '•'•'•    '•'••• 

^  ;^S  il  :  :  :|  iSRfe    ::        :g^    :  :g^ 

|S||^||||||||^|B|  ||  i  ||l  mi 

CATHOBE-RAY-TUBE  DISPLAYS  15-49 


24.  CATHODE-RAY-TUBE  DISPLAYS 

By  T.  Seller 

A  cathode-ray-tube  display  is  a  means  of  presenting  information  concerning  a  signal 
(which,  may  be  any  quantity  that  can  be  put  into  the  form  of  an  electrical  voltage)  in 
terms  of  one  or  more  independent  variables  on  which  the  signal  depends.  Usually  at  least 
one  of  these  variables  is  an  explicit  or  implicit  function  of  time.  The  variables  are  repre- 
sented by  displacements  along  cartesian  or  polar  coordinate  axes;  the  displacements  are 
usually  periodic,  a  single  traversal  along  any  axis  being  called  a  "sweep." 

A  display  is  said  to  be  deflection-modulated  if  the  signal  produces  a  lateral  displacement 
of  the  electron  beam  from  some  base  line  or  sweep;  it  is  said  to  be  intensity-modulated  if 
the  signal  voltage  is  used  to  increase  or  decrease  the  intensity  of  the  electron  beam.  A  de- 
flection-modulated display  is  used  when  quantitative  information  regarding  the  nature  of 
the  signal  is  required,  for  an  intensity-modulated  display  can  at  best  yield  only  qualitative 
information  regarding  the  signal  strength  or  shape.  An  intensity-modulated  display  offers 
the  advantage  of  being  able  to  present  the  signal  in  terms  of  two  variables,  rather  than 
the  single  variable  (usually  time)  of  the  deflection-modulated  display.  Well-known  ex- 
amples of  these  two  general  types  of  display  are  the  usual  form  of  cathode-ray  oscillograph, 
(deflection-modulated)  and  the  television  display  (intensity-modulated). 

Many  different  types  of  display  were  developed  in  connection  with  the  applications  of 
radar  during  World  War  II.  In  many  of  these  an  attempt  was  made  to  present  on  the  two- 
dimensional  surface  of  the  CRT  screen  information  regarding  the  signal  involving  three 
variables,  for  example,  range,  azimuth,  and  elevation.  The  designation  and  geometry  of 
these  various  display  types  are  shown  in  the  following  pages.* 

*  For  a  more  complete  discussion  of  the  subject,  see:  M.I.T.  Radiation  Laboratory  Series,  Vol.  22, 
Cathode  Ray  Tube  Displays,  McGraw-Hill  Book  Co.,  New  York,  1947. 


15-50 


ELECTKO-OPTICAL  DEVICES 


Lkieartfrae  base 


Type  A 

A  deflection-modulated  display,  commonly  used  with  electro- 
static-deflection types  of  cathode-ray  tubes.  Consists  of  a  linear 
horizontal  time  base,  with  the  signal  voltage  applied  to  give  a 
vertical  deflection.  Used  primarily  to  measure  the  time  of 
occurrence  (in  radar,  the  range)  of  the  signal,  as  well  as  its 
shape  and  strength.  There  are  many  modifications  of  this  basic 
display  type  (see  types  K,  L,  Mt  N,  O,  R). 


TypeB 

An  intensity-modulated  (radar)  display  in  which  the  horizon- 
tal sweep  is  synchronized  with  the  antenna  direction,  and  the 
vertical  sweep  is  a  linear  time  base  repeated  with  each  trans- 
mitted pulse.  The  received  signal  is  applied  to  the  control  grid 
or  to  the  cathode  of  the  CRT  in  such  a  polarity  as  to  brighten 
the  screen.  The  display  has  a  large  distortion  from  a  true  map 
when  the  range  sweep  starts  with  the  transmitted  pulse  and  a 
large  azimuth  sector  is  displayed.  By  delaying  the  range  sweep, 
and  by  simultaneous  proper  control  of  the  azimuth  expansion, 
a  good  approximation  to  a  true  map  may  be  obtained,  over  a 
limited  region.  Such  variations  are  called  "delayed  B-scans" 
or  "micro-B  scans." 


TypeC 

An  intensity-modulated  display,  with  azimuth  as  the  horizon- 
tal coordinate  and  elevation  as  the  vertical.  This  is  the  tele- 
vision type  of  scan.  In  radar,  principally  used  in  aircraft  for 
location  of  other  aircraft.  Gives  no  indication  of  range.  This 
type  of  indication  is  very  bad  from  the  signal-to-noise  ratio 
standpoint,  unless  "range-gating"  of  the  signal  (from  informa- 
tion obtained  from  a  type  B  display,  for  example)  is  possible. 
Number  of  horizontal  scanning  lines  used  in  radar  is  quite  small, 
and  signal  appears  on  several  scans  because  of  wide  angle  of 
transmitted  beam. 


CATHODE-RAY-TUBE  DISPLAYS 


15-51 


TypeD 

A  modification  of  type  C  indication,  for  airborne  radar,  giving 
crude  elevation  information  (widely  separated  elevation  lines), 
but  also  yielding  range  informal  ion.  As  any  line  is  scanned 
slowly  from  left  to  right,  a  short  range  sweep  extending  from 
one  elevation  line  to  the  next  is  simultaneously  applied.  Thus, 
for  the  signal  shown,  the  pertinent  data  are:  azimuth,  30°  to 
right;  elevation,  -j-  10°;  range,  2  miles  (assuming  distance  be- 
tween elevation  lines  is  equal  to  3  milesk 

Listed  for  record  purposes  only;  never  used  in  production. 


Range 


TypeE 

Similar  to  type  B,  but  with  range  as  the  horizontal  coordinate, 
and  elevation  angle  as  the  vertical.  Presents  a  distorted  vertical 
cross-section,  since  lines  of  constant  height  are  curved,  as  indi- 
cated. ChieSy  used  where  elevation  angle  (and  not  height)  is 
of  importance. 


Down 


TypeF 

Sometimes  called  a  "spot"  error  display.  No  sweep  is  em- 
ployed, but  the  azimuth  error  of  the  signal  is  indicated  by  the 
horizontal  coordinate  and  the  elevation  error  as  the  vertical. 


Azimuth  error 


15-52 


ELECTRO-OPTICAL  DEVICES 


Inverse  range  Indicated 
by  length  of  "wings0 


Down 


Left 


Right 


Type  G 

Sometimes  called  a  "spot  with  wings"  display.  Similar  to 
type  F,  with  spot  extended  to  a  line  whose  length  is  inversely 
proportional  to  range  to  the  target.  Used  mainly  for  gun-laying, 
the  correct  direction  and  range  being  indicated  when  the  signal 
just  fits  between  the  two  short  vertical  reference  lines. 


TypeH 

Known  also  as  the  "double  dot"  display,  indicating  range, 
bearing,  and  elevation-  A  modified  type  B  display,  in  which 
alternate  range  sweeps  are  displaced  slightly  horizontally,  thus 
producing  two  "dots"  from  a  single  echo.  The  position  of  the 
left  dot  is  that  which  would  be  indicated  by  the  type  B  display. 
The  right  dot  is  displaced  vertically  by  an  amount  such  that  the 
angle  9  is  roughly  proportional  to  the  elevation  of  the  target. 


left-* 


Right 


Typel 

An  intensity-modulated  radar  display,  sometimes  called  a 
"broken  circle"  display,  used  with  a  conical  scanning  antenna. 
Each  echo  is  represented  by  an  arc  of  a  circle  whose  radius  rep- 
resents range,  and  for  which  the  shortness  of  the  arc  indicates 
the  error  in  pointing  at  the  target.  Thus  the  arc  A  represents 
a  target  far  to  the  left  and  below,  while  .#  is  a  target  just  a  little 
above  and  to  the  right.  A  target  dead  ahead  would  be  repre- 
sented by  a  complete  circle. 


CATHODE-RAY-TUBE  DISPLAYS 


15-53 


Synchronizing  0 
- 


Slgnal 


TypeJ 

Essentially  a  type  A  display  bent  into  a  circle.  Useful  for 
accurate  time  or  range  measurement  when  a  continuously  run- 
ning (crystal  or  other)  oscillator  can  be  used.  The  circular 
trace  is  obtained  by  applying  two  sinusoids  from  the  same  source 
to  the  two  pairs  of  deflection  plates,  one  sinusoid  being  90°  out 
of  phase  with  the  other.  The  synchronizing  pulse  which  occurs 
at  0°  also  initiates  the  phenomenon  to  be  observed,  and  the  time 
to  be  measured,  for  example  the  range  of  a  radar  echo  or  the 
delay  in  a  network  is  obtained  from  the  angular  position  of  the 
observed  signal.  The  radial  deflection  is  obtained  by  applying 
the  signal  voltage  to  a  central  electrode  of  the  CRT  (type  3DP1). 
The  sweep  may  be  delayed  so  that  zero  time  occurs  at  0°  on 
some  previous  rotation  of  the  sweep. 


TypeK 

A  radar  display  used  with  systems  sending  out  two  beams  in 
slightly  different  directions.  The  type  A  sweeps  from  the  two 
lobes  are  displaced  slightly  horizontally.  By  adjusting  the 
angular  position  of  the  transmitting  antennas  until  the  signal 
height  of  the  two  corresponding  displaced  signals  is  the  same, 
bearing  or  elevation  information  as  well  as  range  may  be  ob- 
tained. 


TypeL 

Sometimes  called  *  'back-to^ack' '  A  displays.  Used  with  two- 
lobe  radar  systems,  the  signal  S  would  indicate  a  target  to  the 
left.  Changing  the  direction  of  the  antennas  until  the  opposite 
signals  are  of  equal  strength  gives  bearing  or  elevation  of  target 
as  well  as  range. 


15-54 


ELECTRO-OPTICAL  DEVICES 


Range 


TypeM 

A  modification  of  the  basic  type  A  display,  in  which  a  cali- 
brated variable  delayed  "step"  voltage  is  applied  to  one  of  the 
vertical  deflection  plates,  in  order  to  give  more  accurate  range 
information.  In  practice,  the  step  is  shifted  along  the  display 
until  the  signal  to  be  ranged  comes  just  at  the  edge  of  the  step. 


Type  1ST 

Combines  features  of  types  K  and  M  in  order  to  give  more 
accurate  range  and  bearing  or  elevation  information. 


Range 


Type  O 

Sometimes  designated  as  a  "type  A  with  a  notch"  display. 
Useful  for  accurate  ranging  on  an  echo.  The  horizontal  sweep 
speed  is  increased  for  a  short  time  interval,  the  beginning  of  the 
interval  being  adjustable  and  accurately  calibrated.  The  signal 
under  observation  is  placed  in  the  center  of  this  expanded  por- 
tion of  the  sweep,  whereupon  the  reading  of  the  delay  dial  gives 
the  range  directly. 


Range 


CATHODE-RAT-TUBE  DISPLAYS 


15-55 


Type  P  or  PPI 

The  most  widely  used  intensity-modulated  radar  display, 
called  plan  position  indication  (PPI),  in  which  the  range  of  the 
echo  is  given  by  its  distance  from  the  center  of  the  display,  and 
its  bearing  by  the  angular  position.  The  display  may  be  either 
stabilized,  so  that  north  is  at  0°,  or  unstabilized,  in  which  case 
0°  represents  the  ship's  or  plane's  heading.  To  produce  this 
display,  a  radial  sweep  starting  from  the  center  at  the  instant  of 
the  transmitted  pulse  moves  outward  in  the  direction  in  which 
the  antenna  is  pointing  at  the  moment.  The  rotating  sweep 
may  be  produced  by  a  linear  sawtooth  current  in  a  single-asis 
deflection  coil  mounted  around  the  neck  of  a  magnetic  deflection 
CRT,  the  coil  being  rotated  in  synchronism  with  the  antenna. 
Alternatively,  a  stationary  two-axis  coil  may  be  used,  with  x- 
and  2/-axis  sawtooth  currents  whose  amplitudes  are  at  any 
given  time  proportional  to  cos  S  and  to  sin  6  respectively, 
6  being  the  direction  in  which  the  antenna  is  pointing.  Electro- 
static deflection  CRT  may  also  be  used,  sawtooth  voltages 
similarly  modulated  being  applied  to  the  two  pairs  of  deflection 
plates. 

Two  modifications  of  the  centered  PPI  are:  (a)  the  "open- 
center  PPI,"  in  which  increased  accuracy  in  determining  azi- 
muth at  close  ranges  can  be  obtained  by  starting  the  radial 
sweeps  from  a  circle  of  arbitrary  radius,  and  (b)  the  "delayed 
PPI,"  in  which  increased  accuracy  in  determining  the  range  of 
distant  objects  results  from  delaying  the  beginning  of  each 
sweep  from  the  center  by  an  arbitrary  known  time. 


Off-center  PPI  or  Sector  Display 

The  off-center  PPI  is  a  variation  of  the  centered  PPI  described 
above,  in  which  the  center  of  the  pattern  may  be  arbitrarily 
displaced  in  any  direction,  and  the  sweep  expanded,  so  that  any 
desired  portion  of  the  entire  area  within  the  range  of  the  radar 
set  can  be  made  to  fill  the  entire  screen  of  the  CRT.  Such  a 
display  affords  the  possibility  of  greatly  increased  resolution 
while  still  preserving  the  undistorted  map  feature  of  the  PPI. 
This  is  accomplished  in  the  case  of  a  rotating  coil  PPI  by  the 
superposition  of  a  steady  deflecting  field  upon  the  rotating  sweep 
field.  In  the  fixed  x-  and  2/-axis  coil  system,  the  same  result  may 
be  attained  by  proper  control  of  the  delay  and  sweep  speeds  of 
the  two  components.  The  resulting  display  is  often  called  a 
"sector  display,"  although  it  may  not  be  a  complete  sector  if 
the  origin  of  the  pattern  is  not  on  the  CRT  screen. 


15-56 


ELECTRO-OPTICAL  DEVICES 


TypeR 

The  designation  R  (for  range)  applies  to  a  type  A  display  in 
which  a  fast  linear  sweep  is  employed,  the  start  of  the  sweep 
being  delayed  by  a  precision  calibrated  delay  circuit.  Precision 
electronic  time  markers  may  also  be  used.  The  R  display  is 
generally  used  hi  conjunction  with  a  standard  type  A  display  in 
which  the  time  interval  covered  by  the  R  sweep  is  indicated  on 
the  A  sweep  by  intensifying  the  corresponding  time  interval. 
Both  sweeps  may  be  displayed  on  a  single  CRT  at  the  same 
time  by  using  electronic  switching  of  the  two  sweeps  and  vertical 
displacement  of  the  two  traces. 


Range 


Type  RHI 

This  display  is  used  to  indicate  the  range  and  height  of  air- 
craft, hence  the  symbol  RHI,  Since  the  vertical  distance 
(height)  to  be  measured  is  in  this  case  usually  much  less  than 
the  horizontal  distance  (range),  the  useful  area  of  the  CRT  face 
can  be  greatly  increased  by  expanding  the  vertical  dimension. 
The  display  is  usually  obtained  by  applying  simultaneous  saw- 
tooth currents  to  the  x-  and  y-deflection  coils  of  the  CRT,  such 
that  ix  =  A  cos  8-t  and  iy  =  nA  sin  d-t,  where  0  is  the  angle  of 
elevation  of  the  antenna,  and  n  is  the  vertical  expansion  factor, 
usually  between  5  and  10.  In  practice,  the  range  sweep  is 
usually  simplified  to  ix  —  A-t,  the  resulting  distortion  being 
unimportant.  Lines  of  constant  height  are  straight  and  hori- 
zontal. The  range  sweep  may  be  delayed  for  increased  range 
resolution  at  large  distances. 


SECTION  16 
SOUND-REPRODUCTION  SYSTEMS 


AUDIO  FACILITIES  FOR  SOUND  SYSTEMS 
j^^  BY  HOWARD  A.  CHXNN  PAGE 

1.  Sound  Studios 02 

2.  Microphones 04 

3.  Audio  Amplifiers  and  Control  Equipment  06 

4.  Monitoring  Facilities 09 

ELECTROACOUST1C  EQUIPMENT 
BY  R.  J.  KOWALSKI 

5.  Auditorium  Acoustics 11 

6.  Loudspeakers 13 

7.  Amplifiers  and  Control  Equipment 14 

PUBLIC-ADDRESS  SYSTEMS 
BY  R.  J.  KOWALSKI 

8.  Indoor  Sound-reinforcing  Systems 15 

9.  Outdoor  Sound-reinforcing  Systems.  ...  15 
10.  Paging  Systems 16 


SOUND  RECORDING  AND  PROJECTION 

ART.  PAGE 

11.  Recording  Practices,  by  O.  B.  GUNBY.  . .      19 

12.  Projection  Practices,  by  J,  D.  PHYFB.  , .     21 

RADIO  TELEPHONE  BROADCASTING 
BY  HOWARD  A.  CHTNN 

13.  Program  Distribution  Systems 27 

14.  Program  Lines 27 

15.  Broadcasting  Transmitter  Plant , .     28 

16.  Broadcast  Frequency  Allocation , .     30 

17.  Broadcasting  Station  Service 32 

18.  Fidelity     Requirements     of    Broadcast 

System 33 

POLICE  RADIO 
BY  H.  F.  MICEEL 

19.  Frequencies 36 

20.  Power  and  Range 37 

21.  Equipment , 37 


16-01 


SOUND-REPRODUCTION  SYSTEMS 

Sound  energy  Is  transformed  into  some  other  form  and  retransformed  back  into  sound 
for  one  of  three  general  purposes:  it  may  be  desired  to  reproduce  the  sound  instantly  but 
at  some  other  location,  as  in  telephony  or  broadcasting;  it  may  be  desired  to  reproduce  the 
sound  at  some  subsequent  time,  as  in  phonographs  and  sound  pictures;  or  it  may  be  de- 
sired to  reproduce  the  sound  instantly  and  at  the  same  location,  but  with  increased  energy 
content,  as  in  public-address  systems.  The  discussion  herein  is  limited  to  reproduction  by 
electric  means,  or  at  least  partially  by  electric  means,  although  other  means  are  some- 
times used  (e.g.,  acoustic  phonographs). 

An  electrical  sound-reproduction  system  consists  always  of  a  microphone,  or  device  to 
translate  sound  energy  into  electric  energy;  an  amplifier  and  associated  control  equipment, 
to  control  the  loudness  of  the  sound;  and  a  loudspeaker,  or  device  to  translate  electric 
energy  into  sound  energy.  Similar  considerations  govern  the  choice  and  installation  of 
these  devices  in  all  systems. 

In  addition,  in  phonographs  and  sound  pictures  there  are  devices  to  record  the  sound 
permanently  and  to  effect  the  subsequent  reproduction  of  the  sound,  as  well  as  means  to 
synchronize  the  recording  and  reproducing  machines  accurately.  In  telephone  and  radio 
broadcast  systems  a  transmission  link  is  inserted  in  the  circuit;  also  in  telephony  a  com- 
plicated switching  system  is  included  to  permit  any  subscriber  to  talk  privately  to  any 
other  subscriber.  (See  Section  17). 


AUDIO  FACILITIES  FOR  SOUND  SYSTEMS 

By  Howard  A.  Chinn 

1.  SOUND  STUDIOS 

Sound-reproduction  systems  in  common  use  employ  a  single  channel  for  transmission 
and  for  reproduction.  Regardless  of  how  many  microphones  are  utilized  for  a  given  pick- 
up, their  outputs  are  ultimately  blended  into  one  audio  channel.  This  results  in  a  single- 
channel  system  which  imposes  problems  of  a  special  nature  upon  the  design  of  studios  and 
sets  used  for  sound  broadcasting  or  for  sound  recording. 

When  a  person  listens  to  sound  directly,  the  listening  is  done  with  both  ears,  or  bin- 
aurally.  Binaural  hearing  is  indispensable  for  the  localization  of  sound  and  for  sound 
perspective.  When  listening  to  music  in  a-eoncert  hall,  for  example,  one  can  largely  ignore 
disturbing  noises,  provided  they  originate  in  a  direction  that  is  different  from  that  of  the 
desired  sound  and  are  not  of  unreasonable  intensity.  Furthermore,  it  is  even  possible  to 
concentrate  upon  a  particular  section  of  the  orchestra  (for  example,  the  strings  in  pref- 
erence to  the  brass,  or  vice  versa)  and  to  ignore  sounds  that  may  be  of  no  interest,  or  per- 
haps even  unpleasant. 

In  listening  to  a  single-channel  system,  on  the  other  hand,  nearly  all  sense  of  location  of 
the  sound  is  lost,  as  well  as  its  extension  in  space.  (A  simple  but  effective  demonstration 
of  this  is  to  block  one  ear,  while  conversing  in  a  noisy  room.  The  ability  to  continue  to 
understand  the  speaker  will  be  greatly  impaired,  if  not  entirely  destroyed,  because  of  the 
inability  to  exclude  unwanted  sound.)  This  same  limitation  exists  when  sound  is  trans- 
mitted over  a  single  channel  and  reproduced  by  a  loudspeaker  system.  Even  though  both 
ears  are  used  for  listening  to  the  loudspeaker,  the  sound  comes  from  a  single  source  (multi- 
ple-unit loudspeakers,  such  as  are  sometimes  required  on  wide-range,  high-fidelity  systems, 
do  not  change  the  effect)  and  the  ability  to  discriminate  against  undesirable  sounds  is 
almost  completely  lacking.  Hence,  the  sound  that  is  heard  from  radio  or  television  broad- 
casting, and  in  the  cinema  theater,  is  lacking  in  sound  perspective.  The  design  of  studios 
for  sound-reproduction  purposes  must  be  undertaken  with  these  considerations  in  mind. 

By  combining  acoustically  correct  architecture  with  suitable  microphone  pick-up  and 
blending  techniques,  a  single-channel  sound  system  will  produce  pleasing  results.  How- 
ever, the  acoustical  design  problems  are  considerably  more  severe  than  those  encountered 
in  ordinary  architectural  work.  Since  the  ability  to  localize  sound  is  lacking,  it  is  necessary 
to  remove  all  sources  of  confusion  or  interference  incident  to  monaural  listening.  This 

16-02 


SOUND  STUDIOS 


16-03 


may  be  done  by  (a)  proper  control  of  reverberation,  (&)  the  elimination  of  echoes,  and  (c) 
the  elimination  of  disturbing  noise.  The  first  two  items  involve  problems  in  acoustical 
treatment;  the  last  one,  problems  of  sound  isolation — the  two  are  not  necessarily  related. 

OPTIMUM  REVERBERATION  TIME.  Experience  has  shown  that  the  most  satis- 
factory reverberation  time  for  broadcasting  or  sound-recording  studios  is  less  than  that 
which  is  considered  optimum  for  binaural  listening.  (See  Section  12.)  However,  the 
reduction  should  be  no  more  than  is  required  to  obtain  the  desired  result.  An  excessive 
reduction  in  reverberation  time  will  result  in  the  elimination  of  the  reverberant  char- 
acteristics normally  associated  with  the  type  of  sound  being  reproduced.  For  example, 
the  timbre  or  quality  of  a  church  organ  being  played  in  an  acoustically  dead  room  would 
sound  as  if  it  were  out-of-doors.  The  reverberant  quality  associated  with  church  music 
would  be  completely  missing.  The  proper  amount  of  reverberation  must,  therefore,  be 
retained  to  create  the  desired  psychological  effect. 

The  use  of  studios  that  are  acoustically  too  dead  results  in  another  undesirable  effect. 
Musical  organizations  performing  in  an  environment  of  this  type  are  handicapped  in 
several  ways.  The  performer,  discovering  that  his  instrument  does  not  produce  the  sound 
intensity  to  which  he  is  accustomed,  generally  tries  to  produce  the  normal  amount  of 
sound,  with  the  result  that  he  concludes  the  studio  is  a  "difficult"  one  in  which  to  work. 
In  addition,  the  performer  may  find  that  it  requires  special  attention  to  hear  the  other 
members  of  his  group  in  order  to  keep  in  time  and  in  tune.  Finally,  the  listener  gains  the 
impression  that  the  orchestra  is  a  much  smaller  aggregation  than  it  actually  is. 

The  type  of  the  production  dictates  the  optimum  reverberation  time  for  any  given 
studio.  In  general,  studios  intended  for  speech  or  dramatic  productions  are  less  rever- 
berant than  those  used  for  musical  presentations.  It  is  evident,  therefore,  that  no  one 
optimum  will  cover  all  situations.  Furthermore,  since  maximum  studio  usage  requires 
that  a  given  unit  be  capable  of  accommodating  all  types  of  productions,  some  means  of 
adjusting  the  acoustics  is  not  only  desirable  but  actually  essential  for  full  Bexifaility.  In 
Fig.  1  there  are  shown  recommended  minimum  and  maximum  reverberation  times,  at 


2.0 
to 

g    1.8 

§'•• 


K2 
1.0 


2  0.6 

UJ 

£  0.4 

UJ 

(5  0.2 
<e 


I  1  I 

.    SOUND   STUDIO   REVERBERATION  TIME 

RECOMMENDED   RANGE   AT  IOOO   CPS 

(AS    A  FUNCTION  OF   STUDIO  VOLUME) 


5  10  20 

STUDIO   VOLUME -THOUSANDS   OF   CUBIC  FEET 

FIG.  1.     Recommended  Range  of  Sound  Studio  Reverberation  Time  (at  1000  Cycles) 


too 


1000  cycles,  as  a  function  of  studio  volume.  The  most  reverberant  condition  shown  is  suit- 
able for  symphonic  orchestras,  church  organ  music,  and  other  musical  productions  nor- 
mally associated  with  large  halls;  the  least  reverberant  condition  is  suitable  for  dramatic 
productions  and  musical  programs  where  no  '"room  tone"  is  desired. 

It  is  desirable  that  studios  be  designed  to  provide  a  variation  of  reverberation  time  be- 
tween the  two  limits  shown  in  Fig.  1.  This  amount  of  flexibility  makes  it  possible  to  adapt  a 
studio  for  praeticaDy  any  kind  of  production.  The  advantages  of  being  able  to  do  this  are 
obvious.  The  maximum  load  factor  for  every  unit  i&  assured  without  placing  groups  in 
studios  having  unsuitable  acoustical  characteristics.  This  feature  alone  will  justify  many 
times  over  the  additional  design  problems  that  variable  acoustics  entails.  Since  a  studio 
with  variable  acoustics  can  be  used  for  all  types  of  productions,  fewer  studio  units  are 
needed  in  any  given  group.  The  results  are  not  only  greater  performer  satisfaction  but 
also  a  smaller  initial  investment  and  a  smaller  operating  cost  for  the  studio  group  as  a 
whole. 


16-04  SOUND-REPRODUCTION  SYSTEMS 

VARIATION  OF  REVERBERATION  TIME  WITH  FREQUENCY.  In  addition  to  the 
reverberation  time  at  a  specific  mid-range  frequency,  the  manner  in  which  the  rever- 
beration time  varies  with  frequency  is  important.  Various  investigators  have  discussed 
the  shape  of  the  reverberation  characteristic  from  theoretical,  subjective,  experience,  and 
environment  viewpoints.  Their  findings  are  not  conclusive,  however.  Furthermore, 
practically  all  were  based  upon  binaural  listening  conditions. 

Experience  gained  from  broadcasting  and  recording  studio  applications  seems  to  in- 
dicate that,  except  in  the  larger  studios  (i.e.,  those  of  about  50,000  cu  ft  or  more)  the 
reverberation  time  should  be  essentially  independent  of  frequency  (in  practice,  however, 
the  humidity  often  limits  the  reverberation  time  at  high  frequencies) .  In  the  very  large 
studios  some  increase  in  the  reverberation  time  at  the  low  frequencies  is  often  beneficial 
for  the  type  of  productions  for  which  such  studios  are  used. 

In  a  studio  having  variable  acoustics,  provisions  may  also  be  made  to  obtain  variations 
in  the  shape  of  the  reverberation-frequency  characteristic.  By  properly  locating  various 
types  of  acoustical  material  on  the  absorbing  side  of  adjustable  panels  (or  on  the  wall  back 
of  movable  panels) ,  it  is  possible  to  change  the  shape  of  the  characteristic,  within  limits. 
Variation,  at  the  low  frequencies,  from  a  fiat  characteristic  to  a  rising  or  falling  one  may 
often  prove  useful.  A  change  of  20-30  per  cent  above  a  flat  characteristic  at  100  cycles 
to  a  like  amount  below  should  prove  entirely  adequate. 

SOUND  DECAY  RATE.  The  manner  in  which  sound  decays  in  a  studio  is  as  important 
as  the  reverberation  time,  or  even  of  greater  importance.  In  some  studios,  sound  does 
not  decay  logarithmically  as  geometrical  acoustics  assumes.  Experience  indicates,  how- 
ever, that  a  smooth  logarithmic  decay  of  sound  results  in  the  most  acceptable  type  of  stu- 
dio. A  decay  curve  of  this  nature  insures  the  absence  of  discrete  echoes  and  eliminates 
one  of  the  sources  of  confusion  to  monaural  listening  already  mentioned.  For  good 
acoustics  it  is  generally  considered  desirable  to  have  a  constant  decay  rate  for  at  least  the 
first  30  or  40  db. 

The  desired  type  of  sound  decay  can  be  obtained  by  creating  a  diffuse  distribution  of 
the  sound.  This  can  be  done  in  a  number  of  ways,  such  as  employing  a  random  distribu- 
tion of  the  sound-absorbing  materials,  by  the  use  otf  serrated  walls  and  ceilings,  or  by  sub- 
stituting curvilinear  surfaces  for  flat  ones.  It  can  be  shown,  theoretically,  that  curved 
panels,  as  contrasted  to  fiat  serrated  panels,  increase  the  area  over  which  sound  wave  is 
dispersed.  Likewise,  it  follows,  that  multicurved  surfaces  are  an  improvement  over 
cylindrical  ones.  However,  it  is  not  yet  generally  realized  that  from  a  sound  pickup  view- 
point the  diffusion  of  sound  in  an  enclosure  can  be  carried  too  far.  Experience  indicates 
that  an  entirely  adequate  amount  of  sound  diffusion  can  be  readily  obtained  with  serrated 
flat  surfaces.  Consequently,  there  is  little  need  to  resort  to  the  more  complicated  (from 
a  construction  viewpoint)  curvilinear  surfaces,  except  as  may  be  deemed  desirable  from 
an  esthetic  point  of  view. 

ACOUSTICAL  NOISE  LEVELS.  The  residual  noise  in  studios,  from  all  sources, 
should  be  as  low  as  possible.  However,  there  is  a  lower  limit  below  which  it  is  not  justi- 
fiable to  reduce  the  noise  from  either  a  theoretical  or  economical  viewpoint.  The  max- 
imum, sound  intensities  normally  encountered  in  studios  at  the  normal  microphone  pick- 
up position  (relatively  close  to  the  speaker  for  speech  but  relatively  distant  for  music) 
range  from  about  75  db  (above  the  acoustical  reference  level  of  10  ~~16  watt  per  cm2)  for 
speech  to  95  db  for  music.  The  listener,  on  the  other  hand,  indicates  that  noises  that  are 
50  to  60  db  below  the  signal  are  either  unobjectionable  or  not  detectable.  Taking  these 
considerations  into  account,  together  with  the  economic  ones,  it  is  evident  that  a  residual- 
noise  level  25  db  above  the  reference  level  is  quite  satisfactory.  Sound  isolation  methods 
are  discussed  in  detail  in  Section  12. 

BIBLIOGRAPHY 

Morse,  P.  M.,  and  R.  H.  Bolt,  Sound  Waves  in  Rooms,  Rffo.  Modern  Physics,  April  1944,  p.  69.  (In- 
cludes an  excellent  bibliography.) 

Articles  in  the  Journal  of  the  Acoustical  Society  of  America,  Journal  of  the  Society  of  Motion  Picture 
Engineers. 

2.  MICROPHONES 

The  types  of  microphones  most  commonly  employed  for  program  pick-up  purposes  are: 
condenser,  moving-coil  (or  dynamic),  velocity-actuated  ribbon,  crystal  (piezoelectric), 
and  combination  units.  These  last  microphones  usually  combine  a  velocity-  and  a  pressure- 
acttiated  ribbon  or  velocity-actuated  ribbon  and  a  moving  coil  into  a  single  unit.  The 
phasing  between  the  two  units  of  the  assembly  is  arranged  so  as  to  obtain  a  directional  ef- 


MICROPHONES  16-05 

feet,  such  as  a  unidirectional  or  a  cardioid  pattern.  The  characteristics  of  various  micro- 
phones are  discussed  in  detail  in  Section  13. 

The  output  level  of  wide-range,  high-quality  microphones  is  extremely  low— so  low,  in 
fact,  that  the  signal-to-thermal-noise  ratio  existing  at  the  output  terminals  of  the  micro- 
phone often  determines  the  overall  performance  of  the  system.  The  sound  intensities 
existing  at  the  microphone  position  for  the  usual  type  of  program  productions  is  such  that 
a  net  insertion  gain  of  50  to  60  db  is  usually  required  to  raise  the  output  level  to  0  VUT  the 
standard  reference  level  (see  below).  Still  greater  gain  is  required,  of  course,  to  raise  the 
level  sufficiently  for  transmission  over  program  circuits,  for  operating  loudspeakers  and 
recorders,  or  for  modulating  a  transmitter. 

The  output  impedance  of  condenser  and  crystal  microphones  is  very  high,  and,  as  a 
rule,  these  units  are  operated  directly  into  the  grid  of  a  tube.  On  the  other  hand,  the  out- 
put impedance  of  the  moving-coil,  ribbon,  and  combination  microphones  has  ranged  from 
a  few  ohms  to  several  thousand  ohms,  depending  upon  the  type.  Such  a  variation  greatly 
restricted  the  universal  use  of  various  kinds  of  microphones  with  different  equipment. 
The  situation  has  now  been  recognized,  and  microphones  intended  for  broadcasting  pur- 
poses have  been  standardized  at  150  ohms.  In  most  cases,  a  transformer  must  be  supplied 
as  an  integral  part  of  the  microphone  in  order  to  provide  the  standard  output  impedance. 

MICROPHONE  PLACEMENT.  In  determining  the  proper  placement  of  a  micro- 
phone for  a  given  type  of  program  it  is  necessary  to  take  into  consideration  the  directional 
properties  of  the  particular  microphone  being  used  as  regards  both  its  amplitude  and  its 
frequency  response  characteristics.  Each  type  of  microphone  has  a  directional  character- 
istic peculiar  to  that  type,  and  both  the  horizontal  and  the  vertical  plane  directional  pat- 
tern of  the  microphone  must  be  considered  in  connection  with  its  application  and  place- 
ment. Some  microphones  are  unidirectional,  others  bidirectional,  and  still  others  non- 
directional  in  the  horizontal  plane.  These  properties  are  extremely  important  and  useful 
for  discriminating  against  undesired  sources  of  sound  and  for  obtaining  a  desired  relation 
between  sounds  from  different  sources. 

Figure  2  illustrates  the  placement  of  two  bidirectional  microphones  which,  for  the  pur- 
poses of  illustration,  are  assumed  to  have  a  figure  S  directional  pattern  in  the  horizontal 
plane.  The  performance  being 
picked  up  is  assumed  to  consist  of 
a  chorus,  a  soloist,  and  an  accom- 
panying orchestra.  Microphone 
A  is  used  for  the  chorus  and  soloist 
pick-up,  the  artists  being  grouped 
on  both  sides  of  the  instrument. 
The  orientation  of  this  microphone 
is  such  that  the  null  point  or  direc- 
tion of  minimum  pick-up  is  towards 
the  orchestra.  Microphone  A, 
therefore,  is  actuated  primarily  by 
the  chorus  and  soloist  and  picks  up 
very  little  of  the  orchestra  music. 
Microphone  B,  on  the  other  hand, 
is  so  located  as  to  derive  its  major  p^  2  An  r^t^tion  of  the  Placement  of  Bidirectional 
source  of  energy  from  the  orches-  Microphones  to  Achieve  Control  of  Sound  Pickup 

tra,  and  its  null  point  is  towards  the 

singers.  The  electrical  outputs  of  these  microphones,  one  of  which  consists  primarily  of 
the  voices  and  the  other  of  the  musical  accompaniment,  are  then  properly  combined  or 
"mixed"  so  as  to  obtain  the  desired  balance.  The  technician  responsible  for  this  operation 
has  control  over  the  relative  magnitudes  of  the  singers'  voices  and  the  music.  He  can 
adjust  the  relation  between  the  two  and  also  the  overall  amplitude  to  obtain  any  desired 
result.  Still  other  microphones  may  be  used  for  picking  up  sound  effects,  announcements, 
audience  or  crowd  noise,  etc.  By  the  proper  placement  each  microphone  can  more  or  less 
be  confined  to  the  pick-up  of  its  assigned  source  of  sound. 

Microphones  having  relatively  sharp  directional  characteristics  are  sometimes  used  for 
outdoor  or  long-distance  sound  pick-ups.  These  devices  are  particularly  good  for  con- 
fining the  source  of  pick-up  of  sounds  at  a  large  outdoor  gathering  to  a  particularly  in- 
teresting part  of  the  crowd  such  as  a  cheering  section  or  a  band  of  musicians.  In  tele- 
vision pick-ups  they  provide  a  means  for  keeping  the  microphone  otit  of  the  camera  angfe. 

The  use  of  more  than  one  microphone  for  the  pick-up  of  a  given  performer  or  group  of 
performers  working  as  a  unit  (orchestra,  chorus,  etc.)  is  generally  to  be  avoided.  Serious 
frequency  and  delay  distortion  is  likely  to  result  if  more  than  one  instrument  is  employed 
for  the  pick-up  because  under  these  circumstances  each  microphone  will  be  a  different 


16-06 


SOUND-REPRODUCTION   SYSTEMS 


distance  away  from  a  given  source  of  sound.  As  a  result  the  sound  waves  do  not  reach 
each  microphone  at  the  same  instant,  and  the  combined  outputs  of  the  microphones  will 
result  in  complete  or  partial  reinforcement  or  cancelation,  depending  upon  the  resultant 
phase  relationships.  Furthermore,  the  phase  relation  is  dependent  upon  the  frequency  of 
the  sound  source  and  does  not,  therefore,  remain  a  constant  quantity  for  all  sounds. 

In  practice  it  is  sometimes  advisable,  however,  to  countenance  these  potential  sources 
of  distortion  and  use  more  than  one  microphone  for  the  pick-up  of  a  given  group  of  per- 
formers. For  example,  an  orchestra  may  have  a  string  choir  too  small  for  good  balance. 
Under  such  circumstances,  supplementing  the  main  microphone  by  a  strategically  placed 
and  judiciously  used  secondary7  microphone  may  enhance  the  overall  balance.  It  is  impor- 
tant to  note,  in  an  operation  of  this  type,  that  the  contribution  of  the  additional  micro- 
phones must  always  be  maintained  at  a  low  level. 

The  frequency-response  characteristic  of  some  microphones  varies  with  the  angle  of 
incidence  of  the  sound  wave  upon  the  instrument.  Microphones  that  exhibit  this  char- 
acteristic usually  have  a  decreasing  response  to  the  higher  frequencies  as  the  source  of 
sound  moves  around  from  the  front  to  the  side  of  the  instrument.  In  placing  such  micro- 
phones this  characteristic  must  be  taken  into  consideration.  The  unidirectional  dynamic 
and  the  condenser  microphones  are  of  this  type.  The  velocity,  non-directional  dynamic, 
and  crystal  microphones,  on  the  other  hand,  maintain  the  same  relation  between  the  high 
and  low  frequencies  at  all  angles  of  incidence. 

BIBLIOGRAPHY 

Hopper,  F.  L.,  Characteristics  of  Modern  Microphones  for  Sound  Recording,  J.  Soc.  Motion  Picture 

Eng.,  September  1939,  p.  278. 
Marshall,  II.  N.,  and  W.  R.  Harry,  A  Cardioid  Directional  Microphone,  J.  Soc.  Motion  Picture  Eng., 

September  1939,  p.  254. 
Marshall,  R.  N.,  and  F.  F.  Romanow,  A  Non-directional  Microphone,  Bell  Sys.  Tech.  J.,  July  1936, 

p.  405. 

Olson,  H.  F.,  Elements  of  Acoustical  Engineering.     Van  Nostrand  (1940). 
Olson,  H.  F.,  The  Ribbon  Microphone,  Proc.  I.R.E.,  May  1933,  p.  655. 


ADDITIONAL 
PROGRAM 
SOURCES 


3.  AUDIO  AMPLIFIERS  AND  CONTROL  EQUIPMENT 

A  complete  sound  system  entails,  in  addition  to  microphones  for  converting  sound  en- 
ergy into  electrical  energy,  facilities  for:  (a)  amplifying  the  exceedingly  low  microphone 
output  to  a  usable  level;  (6)  blending,  into  a  balanced  whole,  program  elements  from 

several   channels    (known 
as  "mixing") ;  (c)  adjust- 
ing the  balanced  program 
to  the    desired   transmis- 
sion level  without  altering 
TO  OUTGOING  the   balance;    (d)    aurally 
LINE         and    visually    monitoring 
the  transmission. 

These  elements  are 
found  in  audio  systems 
used  for  broadcasting, 
sound  recording,  and 
public-address  applica- 

LEGEND  tions.    Figure  3  illustrates, 

in  block  diagram  form,  a 
typical  audio  system  in- 
corporating the  elements 
listed  above.  The  com- 
ponents of  the  complete 
system  are  described  below 
in  the  order  in  which  they 
appear  in  the  Hock  diagram. 
PRELIMINARY  AMPLIFIERS.  It  is  common  practice  in  reproduction  systems  to 
have  a  mixer  volume  control  associated  with  each  microphone  or  other  source  of  pick-up. 
As  already  noted,  however,  the  output  level  of  microphones  is  so  low  that  any  attenuation, 
prior  to  amplification  of  the  signal,  would  degrade  the  signal-to-noise  ratio  of  the  system 
(the  cause  of  the  noise  at  this  point  being  largely  thermal).  Therefore,  before  any  volume 
coiatroling  ean  be  done,  it  is  necessary  to  raise  the  level  of  the  signal  well  above  the  ther- 


Aa 
AM 
AP 
As 
LS 
FIG.  3. 


BOOSTER    AMPLIFIER 
MICROPHONE    PRE-AMPLJFIER 
PROGRAM    AMPLIFIER 
MONITOR    AMPLIFIER 
LOUDSPEAKER 


M 

MG 

P 

vc 

VI 


MICROPHONE 

MASTER    GAIN     CONTROL 
ATTENUATION     PAD 
MIXER    GAIN    CONTROL 


VOLUME    INDICATOR 
Simplified  Block  Diagram  of  Typical  Sound  System  Studio 
Audio  Facilities 


AUDIO  AMPLIFIERS  AND  CONTROL  EQUIPMENT      16-07 

mal-noise  level.  In  fact,  one  of  the  cardinal  principles  of  good  audio  system  design  is 
never  to  permit  the  signal  level  to  fall  below  the  value  existing  at  the  output  terminals  of 
the  microphone.  It  is  therefore  common  practice  to  insert  a  preliminary  amplifier  be- 
tween the  microphone  and  the  mixer  volume  control  (or  any  other  circuit  element  such  as 
a  sound-effects  filter,  dialogue  equalizer,  etc.).  By  using  this  arrangement  the  mixer  con- 
trol is  introduced  into  a  circuit  at  a  point  where  the  level  of  the  microphone  output  has 
been  raised  considerably  above  the  thermal-noise  level  of  the  circuit.  As  a  result,  when 
attenuation  is  introduced  for  mixing  purposes,  the  signal-to-noise  ratio  is  not  degraded. 

Preliminary  amplifiers  usually  consist  of  one  or  two  stages  of  amplification  having  an 
overall  gain  in  the  neighborhood  of  30  to  40  db.  The  amplifier  must  be  carefully  designed 
because  any  noise  originating  in  it  will  be  amplified  by  the  following  amplifier  stages.  It 
is  therefore  imperative  that  amplifier  noises  such  as  microphonics  and  hum  background 
be  practically  non-existent  in  the  preliminary  amplifier.  The  input  transformer  of  the 
preamplifier,  if  one  is  used,  is  a  particularly  susceptible  point  for  the  pick-up  of  stray, 
unwanted,  interfering  fields.  This  unit  must  be  very  carefully  shielded  both  electro- 
statically and  electromagnetically. 

Low-impedance  microphones,  such  as  the  dynamic  and  velocity  types,  are  connected 
to  the  input  of  the  first  amplifier  tube  by  means  of  a  suitable  transformer.  However, 
microphones  of  this  type  are  basically  voltage-generating  devices;  consequently  their 
output  impedance  (150  ohms  in  broadcasting  practice)  is  not  matched  to  the  input  im- 
pedance of  the  preliminary  amplifier.  Rather,  the  preliminary  amplifier  is  designed  to 
have  a  high  input  impedance,  thereby  realizing  as  much  of  the  open-circuit  voltage  of  the 
microphone  as  possible.  The  input  transformer  is  normally  designed  with  as  high  a  step- 
up  ratio  as  commensurable  with  the  required  response-frequency  characteristic.  The 
leads  connecting  the  microphone  to  the  preamplifier  input  transformer  may  be  several 
hundred  feet  long  without  seriously  impairing  the  performance  of  the  device. 

The  condenser  and  crystal  types  of  microphones  are  connected  directly  to  the  input  of 
the  amplifier  tube  through  a  suitable  network  of  resistors  and  condensers.  In  this  case 
it  is  desirable  that  the  leads  connecting  the  pick-up  device  to  the  amplifier  tube  be  of  very 
low  electrostatic  capacitance.  (Connecting  cable  capacitance  attenuates  the  micro- 
phone's output  voltage  but  does  not  impair  its  response  as  a  function  of  frequency.)  It 
is  common  practice  to  make  these  leads  very  short  by  building  the  preamplifier  into  the 
mounting  which  houses  the  microphone  head.  In  the  floor-stand  type  of  mounting  this 
amplifier  is  sometimes  built  into  the  base  of  the  stand. 

In  the  design  of  preliminary  amplifiers  care  must  be  exercised  that  the  output  stage  has 
adequate  power-handling  capacity.  Since  no  gain  control  is  ever  inserted  between  the 
microphone  and  the  preliminary  amplifier,  the  input  to  the  amplifier  will  vary  over  wide 
limits.  The  output  level  of  the  usual  microphone  under  the  average  conditions  met  in 
practice  has  already  been  noted.  However,  if  a  microphone  is  placed  close  to  a  musical 
instrument  that  is  being  played  very  loudly  (or  near  another  source  of  loud  sound)  its  out- 
put level  may  be  as  much  as  20  db  higher  than  that  "normally"  obtained.  The  output 
stage  of  preliminary  amplifiers  must  be  capable  of  handling  without  overloading  the  out- 
put level  resulting  from  this  input  level.  The  output  of  the  preamplifier  is  usually  matched 
to  the  mixer  circuit  impedance  by  means  of  a  suitable  output  transformer.  For  broad- 
casting applications  the  standard  output  impedances  are  600  and  150  ohms. 

MIXER  VOLUME  COHTROLS.  A  mixer  circuit  is  an  arrangement  of  volume  controls 
for  combining  into  one  program  channel,  in  any  desired  proportions,  program  elements  of 
several  channels.  The  multiple  microphoneT  transition,  and  fading  effects,  which  con- 
tribute so  much  to  program  continuity,  are  obtained  with  mixer  controls.  For  instance, 
separate  microphones  may  be  used  for  a  soloist  and  for  the  accompanying  orchestra  (Fig. 
2)  and  the  outputs  combined  to  form  a  balanced  whole.  Since  the  gain  of  each  micro- 
phone channel  may  be  regulated  independently  of  the  others,  a  method  of  controlling  the 
balance  between  various  pick-ups  is  provided  which  does  not  impair  the  quality  of  the 
individual  sources  of  sound. 

A  mixer  is  also  used  to  "fade  down"  a  musical  transmission  so  that  announcements  or 
talks  may  be  superimposed.  All  these  effects  contribute  a  degree  of  smoothness  to  a 
program  which  would  otherwise  be  impossible.  Because  of  their  use  these  controls  are 
sometimes  known  as  "faders." 

The  mixer  control  is,  in  effect,  a  variable-resistance  attenuation  pad,  usually  of  a  T 
structure,  having  a  constant  or  nearly  constant  input  and  output  impedance.  The  de- 
vice is  capable  of  providing  attenuation  over  a  range  from  0  to  about  120  db.  The  at- 
tenuation generally  varies  uniformly  with  the  angle  of  rotation  of  the  control  knob  through- 
out the  range  from  0  to  approximately  50  db.  From  this  point  to  maximum  attenuation 
the  increase  is  very  rapid,  occurring  in  perhaps  one-tenth  of  the  full  arc  of  rotation.  For 
broadcasting  applications  the  standard  miser  control  impedances  are  600  or  150  ohms. 


16-08 


SOUND-REPRODUCTION  SYSTEMS 


MIXER  MATCHING  NETWORK.  A  mixer  matching  network  combines  the  output 
of  a  number  of  mixer  controls  into  a  single  channel  while  maintaining  correct  impedance 
relations.  The  matching  network  provides  the  proper  load  impedance  for  the  individual 
mixer  controls  and  also  the  desired  source  impedance  for  the  following  circuit  element. 


FIG.  4a.    Differential  Matching  Network  Having  Like  Input  and  Output  Impedance 

One  of  the  simplest  forms  of  matching  networks  is  shown  in  Fig.  4<z.     In  the  general 
case  of  n  mixer  positions,  the  value  of  the  building-out  resistors,  Ri,  is 

R(n  -  1) 


Where  the  input  and  output  impedance  of  the  network  are  assumed  to  be  alike  and  equal 
to  R. 

The  loss  between  the  output  of  any  given  mixer  control  and  the  matching  network  load  is 

db  loss 


FIG.  46.    Minimum-loss  Differential  Matching  Network.    Input  and  output  impedances  are  not  alike- 

If  the  requirements  of  like  input  and  output  impedance  are  waived,  a  lower  loss  network 
can  be  used,  Fig.  46.     Here,  the  building  resistor  value  R%  is 

R(n  -  1) 
j%i>  — 

n 
The  output  impedance  RQ  is 

_         R(2n  -  1) 

RQ  =          5 

The  loss  of  the  network  is 

db  loss  =  10  logio  (2n  -  1) 

In,  tfcJs  case  the  output  impedance  can  be  restored  to  the  same  value  as  the  input  im- 
pedance (or  to  any  other  value)  by  means  of  a  matching  transformer. 


MONITORING  FACILITIES  16-09 

BOOSTER  AMPLIFIER.  The  amount  of  attenuation  introduced  in  an  audio  system 
by  the  mixer  controls  and  the  associated  matching  network  is  often  so  great  that  ampli- 
fication must  be  supplied  before  further  volume  controlling  (see  Master  Volume  Control, 
next  paragraph)  can  be  effected.  The  amplifier  used  for  this  purpose  is  termed  a  booster 
amplifier.  It  usually  employs  one  or  two  stages  of  amplification,  has  a  gain  of  30  to  40 
db,  and  is  designed  to  operate  from  a  source  impedance  and  into  a  load  impedance  of 
finite  value  (e.g.,  600  and  150  ohms  in  broadcast  service). 

Whether  a  booster  amplifier  is  required  between  the  mixer  circuit  and  the  master  gain 
control  is  determined  by  the  signal  levels  existing  for  normal  input  levels  and  normal  set- 
tings of  the  controls.  Its  use  is  indicated  if,  by  its  omission,  the  signal  would  fall  below 
the  level  existing  at  the  output  terminals  of  the  microphone. 

MASTER  VOLUME  CONTROL.  The  master  volume  control  or  "gain"  control  is 
"master"  over  the  output  of  the  individual  mixer  controls.  After  these  mixer  controls 
are  adjusted  to  obtain  the  proper  relation  between  the  various  parts  of  the  performance 
the  resultant  overall  volume  of  the  program  material  may  be  adjusted  to  the  desired  level 
by  means  of  the  master  gain  control  without  affecting  the  balance  that  exists.  This  de- 
vice also  ^permits  the  properly  balanced  performance  to  be  faded  in  or  out  while  the  same 
relation  is  maintained  between  the  individual  parts  of  the  program.  The  master  gain 
control  is  similar  in  construction  and  operation  to  the  mixer  volume  controls. 

PROGRAM  AMPLIFIER.  The  purpose  of  the  program  amplifier  is  to  bring  the  level 
of  the  studio  output  up  to  the  point  necessary  to  permit  its  being  fed  directly  into  the  audio 
amplifier  stage  of  a  transmitter,  into  the  program  line  connecting  the  studio  with  a  trans- 
mitter, into  a  loudspeaker  amplifier  or  a  recording  system.  At  a  network  key  station, 
the  output  of  the  program  amplifier  is  fed  into  a  bus  for  distribution  to  the  proper  network 
or  networks. 

This  amplifier  follows  the  preamplifiers,  mixers,  and  master  gam  control  to  amplify  the 
output  of  the  microphones  or  other  program  source  further.  Although  one  preamplifier 
is  needed  for  each  microphone  in  use,  only  one  program  amplifier  is  necessary  for  a  given 
studio  channel.  The  amplification  obtainable  in  these  units  is  generally  in  the  vicinity  of 
50-60  db.  The  amplifiers  are  usually  capable  of  an  output  level  of  approximately  250  milli- 
watts without  serious  overloading.  The  input  impedance  of  this  amplifier  is  matched  to 
the  output  impedance  of  the  preceding  master  gain  control  by  means  of  a  suitable  trans- 
former. The  output  impedance  is  similarly  matched  to  the  load  which  follows  this  unit. 

BIBLIOGRAPHY 

Chinn,  H.  A.,  Broadcast  Studio  Audio  Frequency  System  Design,  Proc.  I.R.E.,  February  1939,  p.  83. 
Chinn,  H.  A.,  CBS  Studio  Control-Console  and  Control-room  Design,  Proc.  I.R.E.,  May  1946,  p.  287. 
Monroe,  R.  B.,  and  Palmquist,  C.  A..  Modern  Design  Features  of  CBS  Studio  Audio  Facilities,  Pr&c. 
IJ2.J2.,  June  1948,  p.  778. 

4.  MONITORING  FACILITIES 

AURAL  MONITORING  FACILITIES.  A  monitoring  system  consisting  of  a  suitable 
amplifier  and  loudspeaker  is  a  part  of  every  complete  sound  system.  These  facilities  pro- 
vide a  means  for  those  responsible  for  the  production  of  hearing  the  program  exactly  as 
it  is  being  sent  to  the  transmitter,  the  network,  or  the  recorder.  It  is  essential  that  the 
fidelity  of  the  aural  monitoring  equipment  be  in  keeping  with  the  remainder  of  the  broad- 
casting or  recording  system. 

An  output  of  10  to  25  watts  and  an  amplification  of  approximately  50  db  are  generally 
obtainable  from  the  monitoring  amplifier.  The  input  impedance  of  the  monitoring 
amplifier  is  usually  very  high  in  order  that  it  may  be  bridged  across  the  program  circuit  at 
a  convenient  point  without  affecting  the  level  or  impedance  balance  of  the  circuit  to  any 
appreciable  extent.  The  output  circuit  is  provided  with  a  suitable  transformer  to  effect 
an  impedance  match  with  the  particular  type  of  loudspeaker  or  speakers  being  em- 
ployed. 

Broadcasting  and  sound-recording  control  rooms  often  afford  a  limited  amount  of  space 
for  the  installation  of  a  monitoring  loudspeaker,  placing  several  special  requirements  upon 
the  design  of  the  unit.  For  instance,  the  directional  properties  of  the  loudspeaker  must 
be  such  as  to  provide  uniform  coverage  for  all  occupants  of  the  control  room  who  are  con- 
cerned with  the  production  in  hand.  This  requirement  usually  entails  some  special  ar- 
rangement to  injure  uniform  distribution  of  the  high  frequencies.  Furthermore,  since 
the  loudspeaker  may  be  relatively  close  to  the  listener,  multichannel  loudspeakers  (if 
used)  must  be  especially  arranged  so  that  the  separate  sources  of  sound  cannot  be  dis- 
tinguished as  such. 


18-10  SOUND-REPRODUCTION   SYSTEMS 

VOLUME  INDICATOR  FOR  VISUAL  MONITORING.  A  volume  indicator  is  used 
in  order  to  provide  a  precise,  visual  means  of  determining  the  volume  level  of  the  program 
material  being  transmitted  by  an  audio  system.  The  standard  volume  indicator  consists 
of  a  copper  oxide  rectifier  and  a  d-c  indicating  instrument.  The  characteristics  of  the 
rectifier,  together  with  the  dynamic,  electrical,  and  other  performance  characteristics  of 
the  indicating  instrument,  are  all  carefully  standardized.  When  calibrated  and  used  in 
the  prescribed  way,  the  standard  volume  indicator  gives  an  accurate  indication  of  volume 
level.  This  is  expressed  as  so  many  "vu"  above  (or  below)  reference  volume — the  number 
of  vu  being  numerically  equal  to  the  number  of  decibels  that  the  volume  level  is  above  (or 
below)  reference  volume. 

REFERENCE  VOLUME — VU.  Reference  volume  is  that  level  of  program  which 
causes  the  standard  volume  indicator,  when  calibrated  and  used  in  the  prescribed  way, 
to  read  0  vu.  By  definition,  the  reading  of  the  standard  volume  indicator  shall  be  0  vu 
when  it  is  connected  to  a  600-ohm  resistance  in  which  there  is  flowing  1  milliwatt  of  sine- 
wave  power  or  n  vu  when  the  calibrating  power  is  n  decibels  above  one  milliwatt. 

Reference  volume,  as  applied  to  program  material,  should  not  be  confused  with  the 
single-frequency  power  used  to  calibrate  the  volume  indicator.  Speech  or  program  waves 
that  result  in  a  volume  reading  of  0  vu  have  instantaneous  peaks  of  power  many  times  1 
milliwatt  and  an  average  power  which  is  a  fraction  of  1  milliwatt.  In  other  words,  ref- 
erence volume  is  not  1  milliwatt,  except  in  the  special  case  of  sine-wave  measurements. 

DBM  VS.  VU.  For  steady-state  measurements  a  reading  in  "vu"  denotes  a  specific 
single-frequency  audio  power;  for  dynamic  program  indications  vu  denotes  only  a  'Vol- 
ume" level.  This  dual  meaning  of  vu  is  avoided  by  use  of  the  term  "dbm"  for  all  steady- 
state  measurements.  Using  this  terminology,  a  reading  expressed  in  dbm  indicates  the 
power  level  of  a  steady  single-frequency  signal  where  the  number  of  dbm  is  equal  to  the  num- 
ber of  decibels  above  (or  below)  a  reference  power  of  1  milliwatt.  On  the  other  hand,  a 
reading  in  vu  denotes  a  volume-level  indication  of  a  program  signal.  A  vu  reading  can 
be  made  only  on  a  standard  volume  indicator  (since  the  dynamic  characteristics  are  in- 
volved) whereas  sine-wave  power  level  measured  with  the  standard  volume  indicator  or 
with  any  other  suitable  a-c  instrument  can  be  expressed  in  dbm. 

The  practice  of  expressing  measurement-signal  levels  in  dbm,  and  of  limiting  vu  to  ex- 
pression of  dynamic  volume  levels,  has  certain  advantages.  Thus,  dbm  is  a  unit  of  finite 
audio  power,  whereas  vu  may  be  considered  only  a  unit  of  volume  level  and,  as  mentioned 
above,  has  no  connotation  of  finite  power  level. 

However,  it  is  necessary  to  establish  a  relation  between  the  vu  level  to  be  used  for  pro- 
gram peaking  and  the  dbm  level  to  be  used  for  performance  measurements.  It  has  been 
found  that  on  typical  program  peaks  reaching  a  given  crest  amplitude  the  standard  volume 
indicator  reaches  an  indication  8  to  14  db  below  that  reached  on  a  steady  tone  of  the  same 
crest  amplitude.  To  take  into  account  this  8-  to  14-db  difference  in  response,  it  is  standard 
practice  to  require  that  performance  requirements  be  met  at  a  single-frequency  test-tone 
level  10  db  higher  than  the  normal  program  level.  This  will  reasonably  insure  that  sys- 
tem performance  is  within  standards  under  most  operating  conditions. 

TALK-BACK  EQUIPMENT.  A  "talk-back"  system  is  generally  employed  in  order  to 
provide  a  means  for  communication  between  the  control  room  and  the  studio  during  the 
course  of  rehearsals.  This  equipment  consists  of  a  microphone  in  the  control  room,  a 
suitable  amplifier,  and  a  loudspeaker  in  the  studio.  By  means  of  this  equipment  the 
technician  or  the  production  man  in  the  control  room  may  talk  to  and  direct  the  per- 
formers in  the  studio  during  rehearsals. 

The  talk-back  equipment  is  interlocked  with  the  regular  studio  equipment  so  that  the 
studio  microphones  and  the  control-room  monitor  speaker  are  turned  off  whenever  the 
talk-back  equipment  is  energized  for  use.  This  prevents  the  generation  of  an  acoustic 
feedback  because  of  coupling  between  the  input  and  the  output  circuits  of  the  amplifier 
systems.  The  regular  preamplifiers,  studio  channel  amplifier,  and  monitor  amplifier  as- 
sociated with  the  studio  are  sometimes  employed,  by  suitable  switching  means,  for  the 
talk-back  service.  Inasmuch  as  both  services  do  not  function  simultaneously,  this  ar- 
rangement results  in  an  economical  use  of  the  equipment  already  available.  The  switch- 
ing complications,  however,  often  do  not  justify  this  practice.  Under  these  circumstances 
separate  or  partly  separate  talk-back  equipment  is  used. 

BIBLIOGRAPHY 

CHnn,  H.  A.,  D.  K.  Gannett,  R.  M.  Morris,  The  New  Standard  Volume  Indicator  and  Reference 

Level,  Proc.  J.S.B.,  Jan.  lf  1940;  also  Bell  Sys.  Tech.  J.,  Jan.  1,  1940. 
Chinn,  H.  A.,  DBM  vs.  VU,  Audio  Engineering,  March  1948,  p.  28. 


AUDITORIUM  ACOUSTICS 


16-11 


ELECTROACOUSTIC  EQUIPMENT 

By  R.  J.  KowalsM 


5.  AUDITORIUM  ACOUSTICS 

When  an  auditorium  is  designed  for  use  with  a  sound-reproducing  system  one  of  the 
most  important  design  considerations  is  the  acoustic  characteristics.  (See  also  Section 
12).  The  sound  emanating  from  the  stage  loudspeakers  reaches  the  listener  only  after 
being  influenced  by  the  acoustic  conditions  of  the  auditorium.  In  view  of  the  technical 
perfection  of  modern  sound-reproducing  equipment,  it  is  frequently  these  acoustic  con- 
ditions that  determine  whether  the  sound  reaches  the  ear  of  the  listener  with  all  its  original 
naturalness  and  realism  or  whether  it  is  distorted,  unnatural,  and  wholly  unsatisfactory 
to  the  listener. 

The  most  common  acoustic  defects  encountered  in  auditoriums  are  reverberation,  echo, 
resonance,  poor  distribution  (loud  and  dead  spots),  and  noise. 

Optimum  reverberation  times  for  auditoriums  of  various  sises  are  shown  in  Fig.  1.  It 
will  be  noted  that  as  the  volume  increases  the  allowable  reverberation  time  also  increases. 


Optimum  Reverberation  Period  for  Sound 
Motion  Picture  Theatres  at  512  Cycles 


3000 


10,000  100,000 

Volume  !a  Cxi  Ft 
FIG.  1.    Optimum  Reverberation  Times  for  Auditoriums 


1,000,000 


It  will  also  be  noted  that  the  optimum  reverberation  time  recommended  is  from  10  to  20 
per  cent  below  the  usual  values  for  an  auditorium  that  uses  live  talent  without  a  sound- 
reinforcing  system.  This  is  because  there  is  an  abundance  of  sound  energy  and  hence 
reverberation  is  not  necessary  to  augment  loudness.  In  fact,  some  recent  thinking  favors 
overtreating  an  auditorium  with  sound-absorptive  material  to  eliminate  most  acoustic 
defects  and  then  artificially  introducing  the  proper  amount  of  reverberation  in  the  sound- 
reinforcing  system,  by  means  of  reverberation  or  echo  chambers.  These  can  be  designed 
to  give  any  desired  amount  of  reverberation  and  can  be  quickly  and  cheaply  changed  to 
meet  different  conditions. 

It  will  be  noted  in  Fig.  2  that  at  the  lower  frequencies  longer  reverberation  periods  can 
be  tolerated  while  at  the  higher  frequencies  the  allowable  period  is  somewhat  reduced. 


o  ISO 

CM 

If 

c  120 

> 

o    60 

1    40 
0        3 

I 

1 

s 

1 

1 

111 

1 

1 

s 

s 

Optimum  Reverberation-  Frequency  Characteristic 
In  Per  Cent  of  Reverberation  71n>e  et  512  CPS 

^ 

>s 

^ 

-^ 

] 

•^ 

•«> 

*•« 

-«i. 

i  ! 

1 

i 

"p 

*^4 

i 

i  ;x 
i 

N 

' 

i 

i 

0                         100                                                 1000                                               10,000 
Frequency  In  CPS 

FIG.  2.     Optimum  Reverberation  Time  with  Frequency 

By  referring  to  the  table  of  absorption  coefficients  of  building  materials  in  Section  12  it 
will  be  noted  that  most  materials  have  a  different  coefficient  for  different  frequencies. 
Because  of  this,  it  is  possible  by  proper  selection  of  materials  to  arrive  at  a  design  which 
will  give  optimum  reverberation  time  throughout  the  frequency  range.  Owing  to  the 
extension  of  the  low-frequency  and  the  high-frequency  ranges  of  modern  reproducing 
equipment  great  care  should  be  observed  in  the  choice  of  sound-absorbing  materials  and 
treatment. 


16-12  SOUND-REPRODUCTION   SYSTEMS 

Echo  consists  of  a  delayed  repetition,  sometimes  several  rapid  repetitions,  of  the  original 
sound.  It  is  most  often  encountered  in  large  auditoriums,  particularly  those  with  curved 
ceilings  and  walls  and  other  surfaces  sufficiently  remote  from  the  source  of  original  sound 
to  cause  a  definite  time  interval  between  the  arrival  of  the  original  and  reflected  sounds  at 
the  listener's  position.  A  multiple  or  nutter  echo  (several  distinct  repetitions)  is  often 
caused  by  parallel  walls  with  smooth  hard  surfaces.  With  the  extension  of  the  high- 
frequency  range  of  modern  equipment  the  problem  of  echo  and  sound  concentration  was 
somewhat  intensified  because  the  high  frequencies,  or  high-pitched  notes,  on  account  of 
their  short  wavelengths,  are  more  easily  reflected  by  small,  smooth  surfaces. 

Echoes  have  much  the  same  effect  as  reverberation  in  that  they  tend  to  blur  speech  and 
music.  Echoes  are  eliminated  by  first  localizing  them  and  then  applying  light  drapes  or 
other  sound-absorbing  materials,  or  by  breaking  up  the  regularity  of  the  offending  sur- 
faces by  stepping,  angling,  etc.,  thereby  dispersing  or  scattering  the  sound  striking  them. 
When  it  is  necessary  to  add  sound-absorbing  material  to  correct  for  reverberation  time,  it 
is  best  to  apply  it  first  to  the  rear  wall  of  the  auditorium.  Since  the  speakers  are  directed 
toward  this  wall  it  is  usually  the  worst  offender  for  echoes.  If  the  side  walls  are  parallel, 
the  next  best  place  to  apply  treatment  material  is  on  alternate  panels  of  the  side  walls 
with  the  treated  panels  staggered  so  that  no  two  untreated  surfaces  are  opposite  each 
other.  This  helps  to  eliminate  flutter  echo  between  walls.  Flutter  echo  between  ceiling 
and  floor  can  best  be  avoided  by  specifying  heavy  carpeting  with  padding  in  the  aisles  and 
heavily  upholstered  seats. 

The  phenomenon  of  resonance,  or  the  ability  to  vibrate  best  at  certain  frequencies,  may 
occur  either  in  structures  or  in  the  air  in  rooms.  The  effect  is  a  build-up  or  overemphasis 
of  certain  frequencies.  Structural  resonance  usually  is  not  harmful  unless  the  resonant 
body  is  mechanically  coupled  close  to  the  source  of  sound.  An  offending  object  can  usually 
be  located  quickly  by  connecting  an  audio-frequency  oscillator  to  the  input  of  the  sound- 
reproducing  system  and  varying  the  frequency  until  resonance  is  reached.  By  surveying 
the  room  while  this  frequency  is  maintained,  the  vibrating  object  can  usually  be  located. 
The  vibration,  can  then  be  eliminated  by  changing  the  mounting  or  adding  damping  ma- 
terial. Resonance  in  air  chambers  is  usually  encountered  in  small,  highly  reverberant 
rooms;  it  rarely  is  a  problem  in  the  main  body  of  a  large  auditorium,  though  it  may  give 
trouble  in  smaller  sections  such  as  the  stage,  which  usually  has  hard,  bare,  parallel  walls, 
or  under  a  balcony,  in  alcoves,  or  in  foyers.  To  eliminate  such  resonance  conditions  on 
the  stage,  as  well  as  to  help  reduce  reverberation  on  the  stage,  absorbing  material  should 
be  draped  in  the  region  around  the  loudspeakers. 

Poor  distribution  of  sound  (i.e.,  loud  and  dead  spots  due  to  the  shape  of  the  auditorium) 
can  usually  be  overcome  by  the  proper  type  and  proper  orientation  of  the  loudspeakers. 
This  problem  is  discussed  further  in  article  6. 

Noise  may  be  defined  as  any  unwanted  sound.  Noise  is  undesirable  particularly  because 
it  has  a  masking  and  a  frequency  discriminating  effect  on  the  desirable  sounds  which, 
therefore,  require  added  loudness  or  power  to  override  the  noise  An  auditorium  in  a  noisy 
city  location  should  have  its  outside  walls  insulated  against  the  transmission  of  outside 
noises  into  the  auditorium.  See  Section  12  for  a  full  discussion  of  noise  reduction. 

The  following  recommendations  will  serve  as  a  guide  in  designing  an  auditorium  to  get 
the  best  results  from  a  modern  sound-reproducing  system: 

1.  All  seats  should  be  of  the  heavily  upholstered  type. 

2.  Heavily  padded  carpeting  should  be  used  in  all  aisles  and  corridors. 

3.  The  rear  wall,  being  potentially  the  greatest  source  of  echo,  should  be  lined  with  an 
efficient  type  of  sound-absorbing  material  and/or  sloped  or  otherwise  shaped  to  direct  the 
reflected  sound  to  nearby  audience  or  treated  areas  to  prevent  echo. 

4.  Surfaces  with  concave  curvature  should  be  avoided  as  much  as  possible.     If  such 
surfaces  are  necessary  they  should  be  broken  up  with  smaller  convex  flutes  to  disperse  the 
sound  or  heavily  treated  to  absorb  most  of  the  incident  sound. 

5.  Large  unbroken  surface  areas,  except  when  used  for  beneficial  reflections,  such  as 
reflection  from  the  splayed  ceiling  and  walls  at  the  proscenium,  should  be  avoided. 

6.  Long  narrow  auditoriums,  high  ceilings,  and  excessively  long  and  low  balcony  over- 
hangs should  be  avoided. 

7.  The  cubical  content  of  the  auditorium  should  be  made  as  small  as  possible,  compat- 
ible with  the  seating  capacity  and  architectural  design. 

8.  A  rising  slope  in  the  orchestra  floor  should  be  used  to  give  unobstructed  "sound'1 
lines  as  well  as  "sight"  lines. 

9.  All  auditorium  walls  should  provide  sufficient  sound  insulation  to  prevent  extraneous 
noises  from  entering. 

10.  All  machinery  and  ventilating  noises  should  be  isolated  from  the  auditorium. 


LOUDSPEAKERS  16-13 

POWER  REQUIREMENTS.  In  calculating  the  power  requirements  for  a  sound- 
reproducing  system  there  are  several  factors  to  be  taken  into  consideration.  IB  an  out- 
door installation  the  listener  gets  only  the  direct  sound  from  the  speakers,  while  in  a  small 
room  or  auditorium  the  total  energy-  reaching  the  listener  is  the  sum  of  the  direct  energy 
and  the  reflected  reverberant  energy.  Hence  the  total  absorption  of  the  auditorium  must 
be  known  to  determine  the  acoustic  power  required.  It  is  also  necessary  to  know  what 
sound  pressures  it  is  desired  to  attain  before  calculating  acoustic  power.  For  high- 
fidelity  reproduction  of  both  sound  and  music  it  is  generally  assumed  that  sound  pressures 
on  peaks  and  loud  passages  will  reach  20  dynes  per  sq  cm,  or  100  db  above  the  threshold 
of  audibility.  Finally,  since  we  are  chiefly  interested  in  the  electrical  power  needed  in  the 
reproducing  amplifiers,  it  is  necessary  to  consider  the  conversion  efficiency  of  the  loud- 
speakers and  the  coverage  efficiency,  that  is,  the  percentage  of  the  total  radiated  sound 
energy  actually  distributed  to  the  required  area. 

In  making  calculations  for  any  given  installation,  the  following  formula  is  recom- 
mended: 


where  Sy  =  floor  or  seating  area  in  square  feet. 

EL  =  loudspeaker  efficiency  expressed  in  fractions  thus,  50  per  cent  —  0.50. 
AQ  =  average  absorption  coefficient  of  room. 
ST  —  total  area  of  room  surfaces  in  square  feet. 

RF  —  reflection  coefficient  of  seating  area.    This  equals  1  minus  the  absorption  co- 
efi&eient  RF  =  (1  —  AF). 

The  constant  in  this  formula  is  calculated  to  give  a  sound  pressure  of  100  db.  If  less 
acoustic  power  is  required,  as  in  small  rooms,  the  power  as  calculated  by  the  formula  may 
be  decreased  an  equivalent  number  of  decibels.  For  outdoor  reproduction,  the  absorption 
is  considered  infinite,  hence  the  factor  in  brackets  goes  to  unity,  simplifying  the  equation 
to  P  =  O.OOOQSjp/^L.  Assuming  an  efficiency  of  45  per  cent,  which  is  average  for  a  mod- 
ern directional  loudspeaker  of  the  type  used  in  outdoor  installations,  the  formula  becomes 
P  =  0.002$  F.  This  indicates  that,  in  order  to  develop  sound  pressures  of  100  db  without 
the  benefit  of  reverberation,  using  a  loudspeaker  with  an  overall  efficiency  of  45  per  cent, 
it  is  necessary  to  supply  2  watts  of  electrical  energy  for  each  1000  sq  ft  of  audience  area. 

6.  LOUDSPEAKERS 

Through  the  years,  a  good  many  different  types  of  loudspeakers  have  been  developed 
to  convert  electrical  energy  into  acoustic  energy  (see  Section  13).  Among  them  were  the 
head-phone-type  magnetic-diaphragm  units,  the  condenser  speakers,  the  magnetic-arma- 
ture speakers,  and  the  moving-coil-type  speakers.  In  recent  years  the  moving  ooil  or  so- 
called  dyiiarnie  speakers  have  proved  the  most  efficient,  and  hence  this  is  the  only  type 
now  in  common  use.  The  magnetic  field  that  the  voice  coil  moves  in  is  supplied  by  a  field 
coil  energized  with  direct  current  or  by  a  permanent  magnet.  With  the  new  magnetic 
alloys  like  Alnico  it  is  possible  to  get  flux  densities  in  the  permanent-magnet  type  that  are 
as  great  as  those  obtained  when  using  an  electrically  energized  field.  Hence  the  efficiencies 
of  the  two  types  are  comparable.  The  permanent-magnet  type  is  rapidly  becoming  the 
most  popular,  because,  though  it  is  a  little  more  expensive  to  build,  it  saves  the  cost  of  an 
electrical  field  supply  as  well  as  considerable  additional  wiring,  which  is  quite  a  factor  when 
a  good  many  speaker  units  are  located  at  remote  points.  Aside  from  the  type  of  field, 
dynamic  speakers  are  classified  in  two  general  types:  (1)  the  direct  radiator  type  and  <2) 
the  horn  type.  The  direct  radiator  type  uses  a  large  cone  with  a  relatively  snail-diameter 
voice  coil  for  a  radiator.  This  unit  is  generally  mounted  on  a  flat  baffle  or  in  a  cabinet. 
The  horn-type  unit  has  a  relatively  small  diaphragm  driven  by  a  large-diameter  coil.  This 
unit  is  specifically  designed  to  be  used  with  a  directional  type  of  horn. 

The  direct  radiator  or  cone  type  of  speaker  on  a  flat  baffle  is  generally  used  in  small 
spaces  such  as  hotel  guest  rooms,  school  classrooms,  or  hospital  ward  rooms.  Although 
this  type  of  loudspeaker  is  frequently  considered  non-directional,  it  has  a  definite  distri- 
bution angle,  particularly  at  the  higher  audio  frequencies.  For  a  rough  approximation 
at  the  most  important  frequency  range,  this  angle  may  be  considered  to  be  90°,  that  is, 
45°  off  from  either  side  of  the  central  axis.  Since  the  shape  of  the  cone  is  symmetrical, 
this  distribution  angle  is  the  same  for  both  the  horizontal  and  vertical  planes.  In  de- 
termining the  number  of  speaker  units  to  place  in  a  given  room,  not  only  the  power- 
handling  capacity  but  also  the  coverage  must  be  considered.  A  floor  plan  of  the  room 
may  be  sketched  to  scale,  and  speakers  may  be  located,  and  the  coverage  of  each  plotted. 


16-14  SOUND-REPRODUCTION   SYSTEMS 

If  the  distribution  angle  of  one  speaker  does  not  cover  at  least  75  per  cent  of  the  total  area, 
two  or  more  speakers  should  be  placed  along  the  wall  until  this  much  of  the  area  is  covered. 
All  speakers  must  be  connected  to  operate  in  phase.  When  speakers  of  this  type  are  used 
for  incidental  music  such  as  in  a  hospital  or  restaurant  it  is  preferable  to  increase  the  num- 
ber of  speakers  and  hence  cut  down  the  power  per  speaker  to  get  the  most  uniform  sound 
intensity  throughout  the  room. 

For  larger  installations  in  auditoriums  or  in  unconventionally  shaped  rooms  much  more 
efficient  distribution  of  sound  energy  can  be  obtained  through  the  use  of  horn-type  loud- 
speakers with  directional  horns.  The  more  concentrated  beam  distributes  the  sound  in 
the  desired  area  and  at  the  same  time  keeps  it  off  walls  and  ceilings,  hence  helping  to 
avoid  serious  echo  effects.  In  de  luxe  installations  where  high-fidelity  reproduction  of 
music  is  desired  best  performance  can  be  obtained  by  using  a  two-way  speaker  system  with 
multicellular  exponential  high-frequency  horns  and  a  folded  baffle  for  the  low-frequency 
speakers. 

In  auditoriums  or  theaters,  the  location  of  the  speakers  for  best  illusion  is  usually  above 
and  to  the  front  of  the  proscenium  arch.  If  this  location  will  not  allow  for  the  projection 
of  sound  into  the  rear  of  the  orchestra  floor  under  the  balcony,  it  will  be  necessary  to  place 
additional  speakers  to  the  sides  of  the  proscenium  directed  to  cover  this  area.  These 
speakers  should  be  kept  as  high  as  possible  and  should  be  angled  to  get  as  little  sound  as 
possible  into  the  front  rows  of  seats.  When  locating  speakers,  caution  should  be  used  to 
see  that  the  projected  sound  beam  does  not  pass  through  the  microphone  pick-up  area 
where  it  would  cause  acoustic  feedback  and  hence  limit  the  gain  of  the  reinforcing 
system. 

When  more  than  a  single  speaker  unit  is  used,  it  is  essential  that  all  units  operate  in 
phase.  On  better  speakers,  the  phasing  is  carefully  controlled  in  production  and  the 
terminals  of  the  speaker  are  marked  so  it  is  only  necessary  to  connect  all  similar  terminals 
together  to  get  proper  phasing.  If  the  speakers  are  not  marked  it  is  possible  to  energize 
the  speaker  fields  and  then  apply  a  small  d-c  voltage  to  each  voice  coil.  By  reversing  the 
polarity  of  the  battery  it  is  possible  to  determine  to  which  terminal  of  the  speaker  the  pos- 
itive terminal  of  the  battery  should  be  connected  to  give  a  forward  deflection  of  the  speaker 
diaphragm.  If  these  positive  terminals  are  connected  all  the  speakers  will  be  properly 
phased. 

7.  AMPLIFIERS  AND  CONTROL  EQUIPMENT 

Before  selecting  the  proper  amplifier  for  a  given  installation  it  is  necessary  to  determine 
the  power  requirements  as  described  in  article  5.  Strict  adherence  to  this  formula  in  small 
auditoriums  might  indicate  relatively  low  power  requirements;  however,  the  Research 
Council  of  the  Academy  of  Motion  Picture  Arts  and  Sciences  recommends  a  minimum  of 
10  watts  of  audio  power  for  the  smallest  of  theaters.  Though  the  average  power  used  in 
normal  reproduction  will  be  considerably  below  this  figure,  the  reserve  power  will  make 
it  possible  to  reproduce  peaks  and  loud  passages  without  distortion.  This  minimum  of 
10  watts  is  recommended  for  auditoriums  up  to  500  seats.  The  recommended  power  rises 
approximately  10  watts  for  each  additional  500  seats.  In  multiroomed  installations  like 
hotels,  the  total  power  requirements  are  found  by  adding  up  the  power  requirements  of 
the  individual  rooms.  Systems  of  this  type  frequently  have  a  varying  use  factor  for  each 
room  so  that  all  rooms  might  not  have  to  be  supplied  simultaneously.  Although  this 
use  factor  varies  widely  with  different  types  of  installations,  the  suggested  values  for  hotels 
are:  100  per  cent  for  a  single  channel  installation,  90  per  cent  for  two  channels,  75  per  cent 
for  three  channels,  and  60  per  cent  for  four  channels. 

For  theatrical  use  the  amplifiers  should  be  capable  of  reproducing  all  frequencies  within 
the  range  of  50  cycles  to  10,000  cycles. 


PUBLIC-ADDRESS  SYSTEMS 

By  R.  J.  Kowalski 

Public-address  systems  fall  in  two  general  classifications  according  to  application.  A 
system  for  amplifying  speech  or  music  presented  directly  to  a  large  audience,  whether  in 
an  auditorium  or  out  in  the  open,  is  appropriately  named  a  sound-reinforcing  system.  A 
system  by  which  a  speaker  at  a  central  location  addresses  people  simultaneously  at  various 
locations  is  known  as  a  paging  system.  In  general,  the  equipment  specifications  for  sound 
reinforcing,  particularly  indoors,  are  more  exacting  than  those  for  paging  applications. 


OUTDOOR  SOUND-REINFORCING  SYSTEMS  16-15 

8.  INDOOR  SOUND-REmFORCMG  SYSTEMS 

The  essential  requirements  of  a  sound-reinforcing  system  are  that  it  must  pick  up  all 
desired  sounds  and  project  them,  unaltered,  with  sufficient  intensity  and  distribution  to 
all  listeners  within  a  given  area.  For  maximum  effectiveness,  the  system  should  function 
so  as  not  to  detract  the  attention  of  the  listeners  from  the  performers.' 

When  installing  a  reinforcing  system  in  an  auditorium,  particular  attention  should  be 
given  to  the  acoustic  conditions.  A  certain  amount  of  the  sound  energy  emanating  from 
the  loudspeakers  finds  its  way  back  to  the  microphone.  Unless  the  difference  between 
the  level  of  the  sound  energy  leaving  the  speakers  and  that  arriving  at  the  microphone  is 
greater  than  the  gain  of  the  system  amplifier,  the  system  will  oscillate  because  of  acoustic 
feedback.  Directional  microphones  aid  in  avoiding  feedback,  because,  with  them,  it  is 
possible  to  position  the  axis  of  maximum  response  toward  the  desired  sound  and  i;he  null 
axis  toward  the  reflected  sound.  Mounting  the  microphones  in  acoustically  treated  com- 
partments in  the  footlight  trough  also  shields  them  from  considerable  reflected  sound. 
Unless  feedback  can  be  corrected  by  repositioning  the  microphones  or  redirecting  the 
loudspeakers  it  will  be  necessary  to  limit  the  gain  of  the  amplifier  and  hence  the  amount 
of  reinforcing  obtained.  However,  if  the  auditorium  is  well  designed  acoustically,  there 
should  be  no  difficulty  in  getting  sufficient  amplification  before  feedback  occurs. 

The  sound-reinforcing  equipment  in  large  auditoriums  and  theaters  is  usually  custom 
built  and  permanently  installed.  The  microphones  and  preamplifiers  are  broadcast 
quality,  and  the  main  amplifiers  and  loudspeakers  are  generally  of  the  type  used  in  sound- 
motion-picture  reproduction.  The  amplifiers  should  have  uniform  frequency  response 
over  the  complete  range  of  audio  frequencies  from  50  to  10,000  cycles.  The  amplifier 
output  capacity  should  be  such  that  at  least  1  acoustic  watt  per  1000  ft  of  floor  space  can 
be  delivered.  The  amplifier  should  be  able  to  deliver  this  amount  of  power  with  less  tfaaa 
2  per  cent  total  harmonic  distortion  over  the  range  from  50  cycles  to  5000  cycles.  The 
loudspeaker  system  should  have  directional  horns  to  distribute  the  sound  energy  efficiently 
in  the  proper  area.  Many  theaters  employ  two-way  speaker  systems  for  their  reinforcing 
systems  as  well  as  for  the  movie-sound  system.  The  two-way  system  assures  uniform, 
highly  efficient  operation  over  the  complete  audio  spectrum.  The  input  equipment  should 
be  highly  flexible  to  accommodate  a  wide  variety  of  microphone  combinations  because  dif- 
ferent types  of  programs  require  different  pick-up  arrangements.  See  article  2.  Where 
the  pick-up  area  is  great,  such  as  an  entire  stage  area  for  vaudeville  and  stage  productions, 
microphones  are  usually  located  in  the  footlight  troughs,  suspended  from  the  scenery 
drops,  and  sometimes  concealed  in  stage  props.  The  optimum  spacing  of  footlight  micro- 
phones has  been  found  to  be  8  to  10  feet.  In  order  to  control  the  input  from  all  these 
microphones  to  get  the  proper  balance  in  sound  levels,  a  mixer  is  required  with  controls 
for  each  input.  The  most  satisfactory  location  of  the  mixer  console  is  somewhere  in  the 
audience  area,  preferably  at  the  head  of  the  balcony  where  the  operator  can  hear  the 
direct  sound  from  the  system  speakers  and  adjust  the  various  inputs  for  best  balance. 

An  excellent  example  of  a  high-quality  sound-reinforcing  system  is  the  one  at  Radio 
City  Music  Hall,  New  York  City.  Because  of  the  width  of  the  stage,  a  three-channel  re- 
inforcing system  is  used  to  obtain  the  best  possible  illusion.  The  stage  microphones  are 
split  into  three  groups:  right  stage,  center  stage,  and  left  stage.  Each  group  is  fed  through 
its  separate  section  of  the  mixer  console,  then  to  its  separate  amplifier,  and  thence  to  the 
speakers  over  the  proscenium  arch.  The  speakers,  too,  are  split  into  three  groups  which  are 
fed  by  the  three  amplifier  channels.  Therefore,  a  sound  originating  on  the  right  side  of  tiie 
stage  is  picked  up  by  the  right-side  microphones,  amplified  by  the  right  channel  amplifier, 
and  reproduced  by  the  right  group  of  loudspeakers.  As  an  actor  moves  across  the  stage  the 
sound  moves  with  him,  creating  an  excellent  illusion.  Sound  reproduction  in  this  system 
is  so  well  balanced  that,  despite  the  tremendous  size  of  the  auditorium,  the  people  in  the 
remotest  corners  can  hear  as  well  as  those  seated  up  front. 

Besides  these  large,  expensive  systems,  many  small  portable  systems  are  available 
commercially  that  can  be  used  for  meetings,  banquets,  etc.,  in  small  rooms.  These  sys- 
tems generally  contain  microphones,  folding  microphone  stands,  amplifier,  and  loud- 
speakers all  in  one  compact  carrying  case.  The  sides  of  the  case  serve  as  baffles  for  the 
speakers  when  they  are  put  in  use. 

9.  OUTDOOR  SOTJND-REINFORCING  SYSTEMS 

Sound-reinforcing  systems  for  outdoor  use  are  generally  higher  powered  than  indoor 
systems,  because  the  average  area  to  be  covered  is  greater,  and  the  power  per  unit  area  is 


16-16  SOUND-REPRODUCTION   SYSTEMS 

also  greater  since  there  is  no  beneficial  reverberation  to  augment  the  direct  sound.  In 
order  to  utilize  the  available  sound  energy  most  efficiently,  highly  directional  loudspeaker 
horns  are  used  almost  exclusively  for  outdoor  work.  Since  the  low-frequency  cutoff  on 
practical  sized  horns  of  this  type  is  well  up  in  the  audio-frequency  range,  the  system 
fidelity  is  not  generally  as  good  as  that  found  in  indoor  systems.  The  fidelity  could  be 
improved  with  a  more  expensive  speaker  set-up;  but,  in  most  cases,  intelligibility  is  more 
important  in  outdoor  systems  than  fidelity.  Since  the  low-frequency  tones  do  not  con- 
tribute materially  to  intelligibility,  they  may  be  sacrificed  in  the  interest  of  holding  down 
the  practical  size  of  the  equipment.  As  in  indoor  systems,  the  loudspeakers  should  be 
located  to  give  the  best  illusion;  but  they  should  be  carefully  directed  to  prevent  any 
direct  energy  from  reaching  the  microphone  and  causing  feedback.  This  problem  is  ob- 
viously much  less  serious  in  outdoor  than  in  indoor  installations. 

For  indoor  work,  the  ribbon-type  velocity  microphone  is  the  most  popular  because  of 
its  uniform  frequency  response,  its  useful  directional  pattern,  and  its  ability  to  pick  up 
sounds  at  a  considerable  distance.  In  outdoor  work  this  microphone  is  not  too  satis- 
factory because  it  is  somewhat  fragile  and  is  subject  to  extraneous  noises  generated  by  the 
wind  disturbing  the  ribbon.  A  much  better  microphone  for  this  application  is  the  dy- 
namic pressure  type. 

10.  PAGING  SYSTEMS 

Although  the  sound-reinforcing  application  of  public-address  equipment  is  probably 
more  familiar  to  the  layman,  paging  systems  have  become  even  more  important.  During 
the  war  years,  large  permanently  installed  plant  broadcast  systems  became  vital  parts  of 
most  big  factories,  making  it  possible  to  locate  key  men  quickly  in  the  acres  of  floor  area, 
and  to  coordinate  production  activities  throughout  the  plant.  Announce  systems  of 
many  types,  installed  on  practically  all  fighting  ships  of  the  fleet,  proved  to  be  the  most 
important  means  of  internal  communication.  New-type  announcing  systems  for  schools 
make  it  possible  to  communicate  quickly  with  all  rooms  and  to  distribute  educational  pro- 
grams as  desired.  Besides  these,  there  are  hundreds  of  more  commonplace  applications 
like  train  announcers  at  railroad  stations  and  call  systems  in  hospitals,  hotels,  restaurants, 
and  other  business  establishments. 

PLANT  BROADCAST  SYSTEMS.  Though  the  leading  manufacturers  are  building 
equipment  components  specifically  designed  for  industrial  use,  there  is  no  such  thing  as  a 
universal  industrial  sound  system.  The  desired  services,  plant  layout,  and  conditions  of 
operation  vary  so  greatly  that  each  installation  becomes  a  custom-engineered  job.  Some 
of  the  many  services  furnished  by  a  properly  designed  and  managed  industrial  system 
are: 

1.  Paging.     The  telephone  operator  or  a  special  paging  operator,  through  announce- 
ments  to  selected  areas  of  the  plant,  can  quickly  locate  personnel. 

2.  Emergency  and  alarm.     By  using  combinations  of  special  signal  generators  and 
verbal  announcements  the  system  is  highly  effective  in  fire,  damage,  and  accident  control. 

3.  Time  signals,     The  system  may  be  connected  to  the  main  time  clock  to  broadcast 
time  signals  at  preselected  intervals. 

4.  General  announcements.     The  paging  operator  or  one  of  the  plant  executives  may 
use  the  system  to  supplant  the  bulletin  boards  for  messages  pertaining  to  plant  operations. 

5.  Work  music.     To  increase  the  efficiency  of  personnel,  planned  music  programs  may 
be  given  at  periodic  intervals  during  the  work  period.     The  source  of  the  music  may  be 
recordings,  a  wired-in  service,  or  radio  programs. 

6.  Entertainment.     During  rest  or  lunch  periods,  programs  may  be  presented  by  live 
talent,  such  as  employee  groups  or  visiting  celebrities. 

7.  Morale  building.     The  system  may  be  used  to  bring  about  more  personal  contact 
in  personnel  relationship  through  inspirational  messages,  drives,  safety  campaigns,  and 
announcements  of  general  interest. 

In  recent  years  many  scientific  tests  have  been  made  on  the  value  of  music  in  industry. 
These  have  proved  that  periodic  musical  intervals  have  a  beneficial  effect  on  most  types  of 
workers,  but  especially  on  those  engaged  in  repetitive  manual  operations  associated  with 
modern  assembly-line  manufacture.  The  chief  effects  are  the  relief  of  fatigue  and  bore- 
dom and  the  dispelling  of  nervous  tension.  Tangible  results  have  been  accomplished  in 
the  reduction  of  labor  turnover,  reduction  of  accidents,  increased  production,  and  im- 
proved quality  of  product. 

The  central  control  equipment  of  the  average  plant  broadcasting  system  is  located  in  a 
sound-treated  room  that  serves  as  a  studio.  The  central  control  console,  or  mixer,  is 
located  here  to  control  the  various  inputs.  These  inputs  may  include  a  paging  microphone, 
one  or  more  studio  microphones,  one  or  more  executive  microphones  in  the  executive 


PAGING  SYSTEMS  16-17 

offices,  one  or  more  radio  inputs,  one  or  two  phonograph  turntables,  time  clock  signal 
generator,  and  possibly  a  fire  or  other  emergency  alarm  signal  generator.  The  mixer 
console  should  include  an  attenuator  to  control  each  input  and  a  master  attenuator  to 
control  the  overall  level  of  any  combination  of  inputs.  It  should  also  include  a  volume- 
indicator  meter  to  permit  the  operator  to  maintain  the  proper  signal  level  on  the  system. 
The  output  of  the  mixer  is  fed  into  a  program  amplifier  which  in  turn  feeds  a  group  of  zone 
selector  switches  on  the  operator's  console.  In  almost  any  setup  it  is  best  to  divide  the 
plant  into  convenient  zones  or  areas  for  programming  and  paging.  In  a  simple  system  the 
division  might  merely  be  "offices"  and  "factory,"  but  in  a  larger  plant  an  individual  zone 
might  be  established  for  each  building  or  each 'department.  This  makes  possible  selective 
announcements  or  paging  calls  to  any  one  part  of  the  plant.  The  output  of  the  zone 
switches  is  fed  to  the  individual  zone  power  amplifiers  that  drive  the  loudspeakers  in  each 
.zone.  If  the  plant  is  small  and  all  zones  are  in  a  single  building,  h  is  best  to  install  all  the 
zone  power  amplifiers  in  the  studio  or  equipment  room  so  that  they  will  be  in  a  central  loca- 
tion to  facilitate  service  work.  However,  if  the  plant  is  spread  over  considerable  area,  or 
in  many  buildings,  it  is  best  to  mount  the  zone  power  amplifiers  in  the  buildings  they 
serve.  The  input  signal  can  be  fed  from  the  studio  at  the  sone  power  amplifier  at  zero 
level  over  standard  telephone  lines  which  normally  link  most  buildings  of  a  large  plant. 
This  is  more  economical  on  amplifier  power  than  trying  to  feed  high-level  energy  over  long 
lines. 

In  a  larger  system  it  might  be  desirable  to  have  provisions  for  sending  two  different 
programs  to  different  areas  simultaneously.  This  is  especially  helpful  when  an  urgent 
paging  announcement  has  to  be  sent  to  one  zone  during  a  regular  musical  program.  The 
particular  zone  can  be  paged  on  a  separate  channel  without  disturbing  the  music  going 
out  to  the  other  zones.  To  do  this,  it  is  necessary  to  add  a  second  program  amplifier  and 
another  volume  indicator  meter  to  monitor  the  level  on  this  second  channel.  Each  input 
switch  should  be  a  three-position  switch  so  that,  besides  an  "off*  portion,  any  of  the  in- 
puts may  be  connected  to  either  of  the  two  program  amplifiers.  The  soae  selector  switches 
should  also  be  three-position  switches  so  any  zone  may  be  connected  to  either  of  the  two 
program  channels  or  may  be  completely  disconnected.  In  plants  subject  to  serious 
emergencies,  the  switching  arrangement  may  be  so  arranged  that,  whenever  the  alarm 
signal  generator  is  sounded,  it  takes  priority  over  all  other  inputs  and  is  automatically 
fed  to  all  loudspeakers  regardless  of  the  position  of  the  zone  switches.  Alarm  contactors 
to  energize  the  signal  generators  may  be  located  throughout  the  plant  at  critical  points. 

Associated  with  the  studio  equipment  should  be  a  monitor  speaker  with  a  switch  which 
enables  the  system  operator  to  monitor  audibly  any  program  going  out  to  the  plant  or  fco 
check  on  proper  tuning  of  the  radio  receiver  before  connecting  it  to  the  program  bus. 
Adjacent  to  the  studio  should  be  a  suitable  storage  room  for  storing  the  record  library  for 
musical  programs  as  well  as  spare  microphones,  etc.,  for  the  system.  Records  for  in- 
industrial  use  should  be  selected  with  some  care.  In  general  popular  records  are  suitable 
for  factory  working  periods  with  possibly  light  classical  music  for  the  plant  restaurant 
during  the  lunch  hour.  Considerable  research  has  been  done  on  the  psychological  effects 
•of  different  types  of  music.  By  reference  to  the  bibliography  foEowing  this  article,  ad- 
ditional information  on  this  subject  may  be  obtained.  From  a  practical  standpoint, 
since  most  factories  are  rather  noisy  the  recorded  music  selected  should  have  fairly  co®r- 
stant  level.  If  a  recording  having  very  loud  and  very  soft  passages  is  reproduced  in  a 
noisy  area,  the  low  passages  will  be  lost  completely  unless  the  overall  level  is  made  so  high 
that  the  loud  passages  are  annoying. 

The  primary  design  consideration  for  equipment  for  industrial  use  is  roggedneas.  In 
many  cases  the  equipment  will  have  to  operate  continuously  through  a  24-hour  working 
day.  All  components  should  be  conservatively  rated  so  they  are  used  well  witfein  their 
capacity  and  hence  will  give  long  life.  Precautions  should  be  taken  in  the  design  ol  the 
^switching  system  so  that  the  equipment  cannot  be  damaged  by  improper  operation  of  the 
controls.  Occasionally  inexperienced  personnel  will  attempt  to  operate  the  equipment, 
,and  so  these  safeguards  are  necessary. 

The  necessary  overall  fidelity  of  the  equipment  varies  with  the  application  and  tlie 
noise  levels  encountered.  If  the  system  is  to  be  used  solely  for  verbal  annoumoeiaeiita, 
an  overall  frequency  response  of  300  to  3000  cycles  is  adequate;  however,  if  music  is  to  be 
reproduced,  the  response  should  extend  from  50  to  10,000  cycles.  This  wide  freqiaeacy 
range  is  not  merely  to  give  high-fidelity  reproduction  but  also  to  add  needed  definition  to 
music  being  reproduced  under  adverse  conditions.  Noise  generally  occurs  at  specific 
frequencies.  If  the  music  is  reproduced  with  a  limited  band  of  frequencies,  some  of  the 
frequencies  will  coincide  with  those  of  the  noise  and  hence  will  be  masked.  The  loss  ol 
certain  notes  reduces  the  definition  of  the  music  and  makes  it  hard  to  follow.  ^  Of  course, 
.the  difficulty  can  be  overcome  by  making  the  volume  louder  to  override  the  noise,  but  this 


16-18  SOUND-REPRODUCTION  SYSTEMS 

might  make  the  music  annoying.  By  extending  the  frequency  range,  the  definition  may 
be  improved  without  increasing  the  overall  volume. 

Plant  noise  levels  and  the  shape  of  the  building  determine  the  size,  type,  number,  and 
locations  of  loudspeakers.  In  high-noise  areas,  best  results  are  obtained  from  horn-type 
loudspeakers  which  can  direct  the  sound  energy  to  the  important  areas.  However,  if  the 
noise  level  is  below  90  db,  well-baffled  cone-type  loudspeakers  should  be  used.  It  is  pos- 
sible to  get  more  uniform  distribution  over  wide  areas  with  this  type  of  speaker.  The 
power  required  for  any  given  zone  may  be  computed  fairly  accurately  by  means  of  the 
formula  given  in  article  5.  Where  the  existing  noise  level  is  above  100  db,  the  power  as 
computed  by  the  formula  should  be  increased  accordingly.  Most  standard  types  of 
microphones  are  suitable  for  reproduction  from  a  studio ;  but  if  the  microphone  is  located 
in  a  high-noise-level  area  out  in  the  plant,  a  specially  designed  close-talking  microphone 
should  be  used  to  reduce  the  response  to  unwanted  room  noise.  A  phonograph  turntable 
for  industrial  use  should  be  weighted  and  dynamically  balanced,  and  it  should  be  driven 
by  a  heavy-duty  motor  to  insure  constant  speed.  Home-type  record  changers  should 
never  be  used  in  an  application  such  as  this. 

Before  actually  selecting  the  equipment  for  a  particular  plant,  a  thorough  plant  survey 
should  be  made.  This  survey  should  supply  the  following  information : 

1.  Noise.    AH  sources  of  noise  should  be  located  on  a  copy  of  the  plant's  floor  plans  as 
well  as  the  average  level  of  the  noise  in  all  areas.     This  information  can  be  quickly  ob- 
tained with  a  portable  sound-level  meter. 

2.  Coverage.     On  the  basis  of  the  above  measurements,  the  types  of  speakers  should 
be  selected  and  their  locations  marked  on  the  print. 

3.  Plant  zoning.    All  areas  should  be  zoned  with  reference  to  industrial  operations  for 
determining  switching  requirements. 

4.  Locations.    The  location  of  the  studio,  input  sources,  amplifier  equipment,  and  con- 
trols should  be  determined  and  marked  on  the  prints. 

5.  Special  considerations.     Any  special  information  such  as  desired  priority  of  signals 
and  temporary  microphone  locations  should  be  noted. 

On  the  basis  of  the  above  information  it  will  be  possible  to  engineer  a  system  that  will 
exactly  fill  the  needs  of  the  plant.  To  fill  the  needs  of  a  wide  variety  of  different  plants, 
some  manufacturers  have  designed  a  series  of  very  flexible  sectionalized  units  which  may 
be  put  together  like  building  blocks  to  make  up  any  desired  type  of  system  out  of  standard 
equipment. 

NAVY  ANNOUNCE  EQUIPMENT.  A  modern  fighting  ship  is  a  complex  organization 
with  a  huge  staff  of  personnel  engaged  in  a  wide  variety  of  activities  necessary  to  operate 
it.  To  coordinate  all  these  activities  it  is  necessary  to  use  a  shipwide,  selective,  announce 
system.  When  the  ship  is  under  way,  the  orders  originate  on  the  bridge;  when  the  ship 
is  in  port,  the  ship's  control  center  is  the  quarter  deck,  hence  orders  originate  from  this 
station.  The  ship  is  divided  into  sections  so  that  orders  may  be  issued  to  selected  sec- 
tions. In  addition,  each  section  is  subdivided  into  smaller  subgroups  which  can  be  sep- 
arately disconnected  in  the  event  of  battle  damage;  then  if  one  speaker  line  is  shorted,, 
it  may  be  cut  loose  so  that  it  does  not  affect  the  rest  of  the  system.  Besides  the  shipwide 
general  announce  system,  a  large  ship  has  many  more  specialized  systems  such  as:  (1)  the 
engineer's  announce  system  between  the  engineering  log  room  and  the  various  fire  rooms, 
boiler  rooms,  and  other  engineering  spaces;  (2)  the  damage  control  system  between  the 
damage  control  office  and  several  damage  control  stations;  (3)  turret  control  systems  for 
communication  between  the  turret  captain  in  each  of  the  primary  battery  turrets  and  the 
gun  pointer,  the  gun  layer,  and  the  various  shell-handling  and  power-handling  decks  below 
the  turret;  and  (4)  the  secondary  battery  announce  system  communicating  between  the 
gun  directors  and  the  gun  mounts  of  the  secondary  batteries.  Moreover,  there  are  high- 
powered  systems  using  a  single  loudspeaker  that  can  be  directed  for  ship-to-ship,  ship- 
to-dock,  and  ship-to-plane  sound  communication,  and  a  variety  of  low-powered  inter- 
communication systems  between  specialized  points. 

The  requirements  for  ruggedness  in  ship  equipment  are  even  more  exacting  than  in 
equipment  for  industrial  uses.  All  ship  equipment  must  be  shockproof,  and  loudspeakers 
on.  the  weather  decks  should  be  weatherproof,  watertight,  and  blastproof  to  stand  the 
pounding  seas  and  the  terrific  concussion  of  gunfire.  Microphones  are  generally  of  the 
close-talking  dynamic  type  to  reduce  the  response  to  undesired  noises. 

SCHOOL  SYSTEMS.  Small  specialized  versions  of  industrial  systems  have  been. 
designed  for  school  use.  The  control  console  is  located  in  the  principal's  office,  where 
announcements  and  programs  originate.  The  control  console  has  a  selector  switch  for 
each  room  so  that  announcements  may  go  to  any  room  or  any  group  of  rooms.  A  special 
connection  with  a  talk-back  amplifier  makes  it  possible  to  use  each  classroom  speaker  as 
a  microphone.  In  this  way,  it  is  possible  to  establish  two-way  communication  between  the 


RECORDING  PRACTICES 


ie-i9 


Available  inputs  besides  microphones  include  a  radio 


principal  and  any  of  the  teachers. 

tuner  and  a  record  player. 

in  J^  f  a^.fe*^es<>f  school  equipment  should  be  simplicity  of  controls  and  relatively 

low  cost,_which  is  achieved  by  a  slight  sacrifice  in  overall  quality  of  reproduction,  while 

maintaining  a  safety  factor  in  the  choice  of  component  parts  to  insure  dependability  of 

operation . 

BIBLIOGRAPHY 

Biims-MeyerH.   Music  iir  Industiy,  Meckamcal  Bngineerins,  January  1943. 

PublSheS"  aS^^  Programming  Music  in  Industry,  AW1  Soc.  Compaq  Ao^ocs,  and 

Halpin,  0  D.,  Industrial  Music  and  Morale,  /.  Actwiticat  Sec.  Am. 
Music  in.  Industry,  Industrial  Recreation  Assodatkai  (1944) 
belvm,  Ben,  Programming  Music  for  Industry,  J.  Acoustical  Soc.  Am.*  October  1943, 


SOUND  RECORDING  AND  PROJECTION 


11.  RECORDING  PRACTICES 

By  O.  B.  Guuby 

Sound  motion  pictures  are  released  extensivery,  either  16  mm  or  35  mrn,  depending  upon 
the  application.  The  use  of  16-mm  film  is  on  the  increase  in  advertising  and  educational 
fields.  The  majority  of  the  studios  make  all  their  pictures  oa  35-rnm  film,  even  though 
some  of  them  will  be  reduced  to  16  mm  before  release,  because  of  the  greater  fiexibiliiy 
obtainable  from  the  35-mm  equipment  commercially  available. 

Recent  trends  in  the  design  and  operation  of  sound-motion-picture  equipment  include 
the  general  use  of  electronic  mixers  (volume  compressors)  or  limiters  to  wsary  the  volume 
range  of  the  recordings  or  to  prevent  overloads  on  loud  signals  Also,  particularly  in  the 
recording  of  music,  synchronously  driven  acetate  recordings  are  often  made  so  that  an 
immediate  playback  of  the  recorded  material  is  possible,  thus  permitting  a  quick  and  ac- 
curate check  on  quality. 

A  block  diagram  of  a  typical  production-type  dialogue  channel  is  shown  in  Fig.  1.  The 
same  type  of  a  channel  may  be  used  for  recording  sound  effects  or  for  the  recording  of 


SOUND  STAGE 


RECORDER  ROOM 


FIG.  1.     Simplified  Scoring  or  Dialogue  Recording  Cfaannel 

music,  which  is  usually  referred  to  as  scoring.  For  scoring,  however,  it  is  customary  to 
use  more  microphone  inputs  than  are  indicated  for  a  dialogue  channel.  Since  the  picture 
is  usually  made  before  the  music  is  recorded,  a  motion-picture  screen  is  generally  provided 
on  a  scoring  stage  so  that  the  orchestra  leader  can  watch  the  picture  while  he  is  directing 
the  orchestra, 


16-20 


SOUND-BEPRODUCTION  SYSTEMS 


In  a  dialogue  channel  the  microphones  are  usually  located  4  ft  or  less  away  from  the 
actors,  but  out  of  range  of  the  camera.  The  microphones  are  usually  mounted  on  long 
adjustable  booms  so  that  they  can  be  moved  to  follow  the  action  closely  and  yet  avoid  the 
camera.  Microphones  of  the  unidirectional  type  are  used  frequently  because  of  their 
ability  to  discriminate  between  the  wanted  sound  from  the  set  and  the  extraneous  noises 
from  other  directions. 

The  sound  stage  in  which  the  recording  channel  is  used  must  be  constructed  to  exclude 
external  noises.  This  frequently  involves  double  wall  construction  with  intervening  air 
spaces  and  accoustic  treatment  on  the  interior  to  provide  the  desired  reverberation  time. 

The  mixer  is  located  near  the  set  in  a  position  where  the  operator  has  an  unobstructed 
view  of  the  action  being  recorded.  This  helps  him  to  anticipate  changes  in  mixer  adjust- 
ment to  suit  the  sound  source.  Monitoring  is  accomplished  with  high-fidelity  headphones. 
The  remainder  of  the  recording  channel  may  be  located  in  a  small  room  on  the  sound 
stage,  in  a  recording  truck  parked  adjacent  to  the  stage,  or  in  a  centrally  located  building 
on  the  studio  grounds  and  connected  to  the  sound  stage  by  suitable  transmission  lines. 

Figure  2  is  a  block  diagram  of  a  typical  rerecording  channel.  It  is  essential  that  the 
console  be  installed  in  a  room  having  acoustic  properties  comparable  to  those  of  an  average 


-DIALOGUE  CHANNEL 


FILM    LL_4MPT|     M«XEI»   UBOOSTHRU VARIABLE |_j«"| 

PHONO.  [-jPBE.AMP.fl         pAp       j-j       AMp         fl         pAD       {-j^l 


FJt-M       M  I]  COMPEN-LJ  BOOSTER  M      M 

[   PHONO.  |-[PRE-AMP.[-|     gATOR     [-[       AHP.      f|        , 


FILM      Up, 

PHONO.  np 


RERECORD1NQ  REVIEW  ROOM 


RECORDER  EQUIPMENT  ROOM 


FIG.  2.     Simplified  Rerecording  Channel 


theater  so  that  the  desired  sound  quality  may  be  determined  through  the  monitor  speaker 
system,  as  the  numerous  sources  are  blended  together  to  make  a  final  sound  track. 

The  various  sound  sources  are  usually  recorded  on  separate  films  in  order  to  provide 
the  desired  degree  of  flexibility.  In  general,  the  sound  tracks  will  include  dialogue,  music, 
and  any  number  of  sound-effect  tracks  that  may  be  required.  These  films  are  threaded 
in  separate  film  phonographs  whose  audio  outputs  are  individually  controlled  by  the 
operator  at  the  mixer  console.  The  threading  of  the  films  is  indicated  by  suitable  marks 
so  that  sound  and  picture  are  synchronized.  The  various  film-handling  machines  are 
driven  by  Selsyn  motors  which  provide  an  electrical  interlock  during  the  operating 
cycle. 

During  a  rehearsal  the  film  phonographs  and  the  projector  are  run  by  the  Selsyn  driving 
system,  and  the  operator  varies  the  audio  signals  from  the  film  phonographs  to  fit  the  mood 
of  the  picture.  Several  rehearsals  are  usually  required  in  obtaining  the  desired  effect. 
Once  this  has  been  achieved,  the  films  are  rethreaded  and  another  run  is  made,  including 
the  film  recording  machine.  During  this  run  the  operator  endeavors  to  repeat  the  changes 
in  level,  etc.,  that  were  made  during  the  successful  rehearsal.  The  procedure  is  repeated, 
if  necessary,  until  a  satisfactory  rerecording  is  made. 

During  the  rerecording  process,  in  addition  to  changes  in  level  that  are  made  in  the 
incoming  signals,  changes  may  also  be  made  in  the  frequency  response  to  adjust  for  day- 
to-day  differences  in  the  original  recording  or  to  obtain  certain  special  effects  such  as  tele- 
phone quality,  old  phonograph  quality,  and  so  forth. 


PROJECTION  PRACTICES 


16-21 


The  more  progressive  studios  usually  have  facilities  for  the  con* ~>I  of  reverberation. 
A  reverberation  chamber  is  one  means  of  control.  A  portion  of  thx  and  requiring  re- 
verberation is  fed  into  a  loudspeaker  in  the  chamber;  the  sound  is  picked  up  by  a  micro- 
phone located  in  the  same  room  and  mixed  with  the  original  sound  to  obtain  the  desired 
effect. 

A  third  type  of  recording  channel,  the  single-film  system,  is  shown  in  Fig.  3.  This 
channel  is  generally  used  in  the  field  for  the  original  recording  of  newsreels  and  to  a  lesser 
extent  for  recordings  on  locations  impractical  for  recording  with  the  more  cumbersome 
production-type  channels.  As  the 
name  implies,  tH«  system  uses  one 
film  on  which  both  the  picture  is 
photographed  and  the  sound  is  re- 
corded. 

The  more  modern  equipments 
use  a  specially  designed  camera 
having  in  it  a  mechanically  filtered 
drum  on  which  the  light  beam  from 
a  compact  recording  optical  system 
can  be  focused.  With  some  equip- 
ments, a  conventional  camera  is 
used  and  the  sound  is  recorded 
directly  on  the  camera  sprocket. 

The  amplifier  is  designed  to  have 
small  size,  light  weight,  and  low 
power  drain,  and  the  facilities  pro- 
vided are  kept  to  a  minimum. 
Power  for  the  amplifier,  exposure 
lamp,  and  camera  motor  are  often 

provided  by   a  single  small  low-  FlG>  3     Sn^  Fdm  R^jrf^  Channel 

voltage  storage  battery. 

Some  newsreel  channels  use  class  B  pushpull  recording  since  this  type  of  track  provides 
excellent  noise  reduction  by  optical  means  and  does  not  add  to  the  size,  weight,  or  power 
dram  of  the  equipment.  This  track  requires  rerecording  to  standard  track  before  it  can 
be  released  to  theaters,  but  since  it  is  always  rerecorded  to  provide  sound  effects,  music, 
etc.,  this  requirement  presents  no  problem  in  the  production  of  the  newsreels. 

Modern  recording  equipments  are  capable  of  producing  films  having  a  fiat  frequency 
response  over  a  range  greater  than  30  to  10,000  cycles  and  a  volume  range  in  excess  of  50 
db.  However,  in  practice  it  is  usually  found  desirable  to  restrict  the  frequency  range  from 
about  70  to  7000  cycles  and  the  volume  range  to  approximately  20  to  25  db.  This  limita- 
tion in  frequency  and  volume  range  is  a  compromise  resulting  from  the  necessity  of  evolving 
a  technique  that  provides  commercially  acceptable  sound  quality  while  recognising  the 
following  variable  factors:  (a)  reasonable  quality  tolerances  for  each  of  the  steps  in  sound 
motion  picture  production,  (6)  the  wide  variety  of  conditions  under  which  sound  films  are 
reproduced  in  the  many  theaters. 


12,  PROJECTION  PRACTICES 

By  J.  D.  Phyfe 

The  sound-reproducing  system  of  a  modern  motion-picture  theater  is  the  combined  prod- 
uct of  many  highly  specialized  arts  and  sciences,  embracing  the  fields  of  optics,  acoustics, 
electronics,  and  mechanics. 

The  following  description  of  the  major  component  items  of  a  typical  sound-reproducing 
system  will  illustrate  how  these  components  are  combined  into  a  complete  system. 

The  equipment  housed  in  the  projection  room  usually  consists  of  two  picture  projectors 
and  associated  lamp  houses,  two  soundheads  which  are  mounted  below  the  projector 
mechanisms,  and  an  amplifier  system.  The  projectors  and  soundheads  for  all  theaters 
are  quite  similar,  the  power  output  rating  of  the  amplifier  and  loudspeaker  systems  being 
modified  to  compensate  for  changes  in  the  seating  capacities  of  various  theaters. 

The  sound-reproducing  industry  has  set  progressively  higher  standards  of  performance, 
as  exemplified  hi  constant  development  work  fostered  by  all  manufacturers  of  theater 
equipment.  Tentative  standards  of  reproduction  have  been  established  by  the  Research 
Council  of  the  Academy  of  Motion  Picture  Arts  and  Sciences. 

The  film,  upon  which  are  photographed  both  the  picture  and  sound  records,  must  be 
moved  through  the  mechanism  of  the  soundhead  at  a  constant  rate  of  speed  to  insure  a 


16-22 


SOUND-BEPKODTJCTION  SYSTEMS 


high  quality  of  sound  reproduction  free  from  "wows"  or  "flutter/'    These  terms  denote 
minute  variations  in.  the  speed  of  the  film. 

Advances  have  been  made  in  the  design  of  the  film-moving  mechanism  to  reduce  to  a 
minimum  irregularities  in  speed  of  the  film.  Two  types  of  film  motion  filter  are  in  cur- 
rent use,  both  employing  a  film-driven  rotating  drum  coupled  to  a  flywheel.  One  type, 


OH  Filrr 


Outer  Case 

Sotfdly  Connected 

to  Drum  Shaft 


Inner  Flywheel 
Free  to  Rotate  on  Shaft. 
Driven  by  Oil  Fiim  Only, 


FIG.  4.    Soundhead  Film  Motion  Filter  (Rotary  Stabilizer) 


known  as  the  Rotary  Stabilizer  (see  Fig.  4),  consists  of  two  flywheels  connected  by  a 
viscous  medium.  The  other  type  utilizes  a  solid  flywheel  in  conjunction  with  a  dampened 
idler  roller  for  its  filtering  action. 

A  picture  of  a  modern  soundhead  is  shown  in  Fig.  5.  The  view  is  of  the  "operating 
side"  of  the  unit  through  which  the  film  passes. 

Light  from  a  source  termed  an  "exciter  lamp"  passes  through  an  optical  system  where 
the  dimensions  of  the  beam  are  rigidly  defined  into  a  narrow  slit  0.00125  in.  wide  and 
0.084  in.  long.  This  slit  image  is  focused  upon  the  sound  track  of  the  film,  which  is  moving 


FIG.  5.    Operating  Side  of  Modern  Soundhead 

and  presenting  a  continuously  variable  ratio  of  clear  film  to  the  dark  or  exposed  area  of  the 
film  that  is  being  scanned  by  the  light  (Fig.  6).  The  film  effectively  serves  to  control  the 
transmission  of  the  light  in  conformity  with  the  light  and  dark  areas  that  comprise  the 
souad  record.  The  variation  in  the  transmission  of  light  is  translated  into  &  corresponding 
variation  in  current  by  means  of  a  photoelectric  cell.  (See  Section  15,  article  7.) 


PROJECTION  PRACTICES 


16-23 


Because  the  photocell  currents  are  weak,  it  is  necessary  to  provide  a  means  of  ampli- 
fication; this  may  consist  of  a  one-  or  two-stage  voltage  amplifier.  Amplifiers  are  usually 
separate  units,  either  incorporated  into  the  soundhead  or  mounted  on  the  front  wall  of  the 
projection  booth  near  the  soundhead.  They  may  also  be  placed  in  the  main  amplifier 
rack  and  coupled  to  the  soundhead  by  means  of  transformers  or  suitable  low-capacity 
coaxial  cables. 

Further  amplification  of  the  photocell  voltage  is  furnished  by  the  main  or  power  ampli- 
fier, raising  the  low-level  currents  to  a  satisfactory  value  where  they  can  be  made  to  operate 
the  theater  loudspeaker  system. 

In  order  that  the  projectionist  be  informed  constantly  of  both  the  volume  level  and 
quality  of  the  sound,  a  small  monitor  loudspeaker  is  installed  in  the  projection  room. 


Pboto-ceU 


•Scanning  Beam  of  Ught 
Sound  Track 


Direction 

of 
Him  Travel 


Photo-cell 


/        Objective  I 
^ —  Condensing  Lets 

FIG.  6.     Optical  System  of  Modern  Soundhead 

Equalizers  are  frequently  employed  to  adjust  the  electrical  response  characteristics  of  the 
amplifier  system  so  as  to  provide  optimum  acoustical  results  in  a  given  theater. 

It  is  common  practice  to  select  the  output  of  the  desired  soundhead  by  alternate  switch- 
ing of  the  exciter  lamp  currents  or  by  selecting  the  audio  output  of  either  soundhe&d  by 
means  of  a  "change-over  switch"  or  '"fader." 

The  projectionist  is  notified  of  the  proper  time  for  making  the  change-over  between 
projection  equipments  by  two  small  cue  marks.  These  marks  will  appear  in  the  picture 
area  of  the  film  and  are  visible  on  the  screen.  The  cues  are  placed  several  seconds  apart 
on  the  film  as  it  passes  through  the  outgoing  projector.  The  first  mark  signals  she  oper- 
ator to  start  the  motor  of  the  incoming  machine.  The  second  cue  mark,  appearing  shortly 
after  the  machine  has  attained  full  operating  speed,  marks  the  point  of  actual  change-over. 
The  picture  is  switched  by  an  electric  dowser  actuated  by  a  foot  switch  operating  in  syn- 
chronization with  the  sound  change-over. 

TWO-WAY  LOUDSPEAKER  SYSTEM.  The  loudspeakers  employed  in  tbeater 
sound  reproduction  are  located  behind  the  picture  screen.  Small  perforations  in  the 
screen,  not  noticeable  from  the  seating  area  of  the  theater,  permit  the  sound  to  pass  readily 
through  the  screen.  A  typical  two-way  loudspeaker  system  is  shown  in  Fig.  7. 


16-24 


SOUND-BEPRODUCTION  SYSTEMS 


The  wide  frequency  range  and  the  large  power-handling  requirement  of  modern  theater 
loudspeaker  systems  cannot  be  met  by  a  single  speaker  mechanism.  The  result  has  been 
the  development  of  the  two-way  loudspeaker  in  almost  universal  use  today.  For  the 
higher  frequencies,  a  unit  having  a  small  light-weight  diaphragm  coupled  to  a  multicellular 


FIG.  7.     Two-way  Theater  Loudspeaker  System 

directional  horn  is  employed.  The  low-frequency  portion  of  the  signal  is  assigned  to  a 
unit  having  a  larger  diaphragm  coupled  to  a  very  large  horn  or  baffle.  The  division  of  the 
high-  and  low-frequency  components  is  accomplished  electrically  through  a  "cross-over 
network."  The  schematic  diagram  of  such  a  network  appears  in  Fig.  8.  The  cross-over 
frequency  is  in  the  region  of  400  cycles. 

Crossover  Network 


.  Hfgh-Frequenqy 
Reproducers 


_  Low-Frequency 
Reproducers 


FIG.  8.    Schematic  Diagram  of  Crossover  Network  for  Two-way  Loudspeaker  System 


A  block  schematic  of  a  complete  theater  sound-reproducing  system  is  shown  in  Fig.  9. 
The  relative  circuit  positions  of  the  components  covered  above  may  be  readily  observed. 

RECENT  DEVELOPMENTS.  Recent  achievements  in  motion-picture-sound  engineer- 
ing are  the  development  of  the  control-track  system  of  reproduction  and  the  drive-in  type 
of  theater.  The  first-named  system  employs  a  control  track  consisting  of  variations  in 
the  area  of  exposure  of  the  small  portion  of  film  lying  between  adjacent  sprocket  holes. 


RADIO   TELEPHONE   BROADCASTING 


16-25 


The  system  has  been  utilized  to  produce  a  96-cycIe  tone,  the  frequency  being  governed  by 
the  number  of  sprocket  holes  per  second  passing  a  small  scanning  %ht.  The  variations 
in  light  actuate  a  separate  photoelectric  cell  as  described  earlier  in  this  chapter.  The 
photocell  current,  after  amplification,  is  rectified  and  used  as  a  control  voltage  to  regulate 
the  output  level  of  the  main  theater  amplifier  and  to  cut  into  operation  an  auxiliary  loud- 
speaker system  placed  at  each  side  of  the  picture  screen.  This  complete  system  permits 
a  tremendous  volume  range  not  otherwise  obtainable  and  provides  a  wider  source  of  sound 


SOUND 

EXC.   LAMP 

SUPPLY 

Hta»-nt£m*£i*e 

SPEAKER 

j-0—  DO 

SWITCH 

EXC.   LAMP 
SUPPLY  VOLTAGE 

1 

SOUNDHEAD 

*  

{ 

1 

PHOTOCELL. 
POLARIZlKa 
VOLIAOE— 

AUDIO: 

PHOTOCELL 
POLARIZING 

VOLTA3E  — 

AUDIO 

Hb-^HDO 

SOUNDHEAD 

2 

-c  

t 

o 

t       SOUND 
HANGE-OVER 
SWITCH 

*     EXC.   LAMP                             MONITOR 

FIG.  9.     Overall  Block  Diagram  of  Theater  Sound-reproducing  System 

during  loud  passages  than  speakers  alone  placed  behind  the  screen;  it  heightens  the  dra- 
matic effect  of  certain  loud  passages, 

DRIVE-IN  THEATERS.  The  popularity  of  the  drive-in  type  of  theater  has  Increased 
considerably  during  the  last  few  years.  The  picture  is  shown  on  a  large  outdoor  screen. 
The  patrons  remain  seated  in  then-  automobiles,  which  are  located  in  an  arc  to  permit 
viewing  the  picture  through  the  windows.  The  trend  is  toward  the  use  of  individual 
loudspeakers  which  may  be  placed  inside  the  cars. 

BIBLIOGRAPHY 

Kimball,  H.  R.,  Applications  of  Electrical  Networks,  J.  Soc.  Motion  Pictwre  Engineers,  Vol.  31  (October 

1938). 
Levinson,  Nat.,  and  L.  T.  Goldsmith,  Vitasound,  J".  Soc.  Motion  Picture  Engineers,  VoL  36  (August 

1941). 
Reiskind,  H.  I.f  Reproducing  Systems,  J.  Soc.  Motion  Picture  Engineers,  VoL  36  (August  1941). 


RADIO  TELEPHONE  BROADCASTING 

By  Howard  A.  Chima 

Radio  broadcasting  is  a  means  for  delivering  intelligence  for  general  reception  at  distant 
points.  A  complete  system  consists  of:  (a)  a  radio  broadcasting  transmitting  system; 
(b)  the  medium  through  which  transmission  takes  place;  (c)  a  number  of  receiving  in- 
stallations. 

A  radio  broadcasting  transmitting  system  consists,  essentially,  of: 

1.  A  studio,  stage,  theater,  auditorium,  or  other  suitable  place  for  the  performance  that 
is  to  be  broadcast.     (See  article  1,  above.) 

2.  An  acoustoelectric  device  (microphone)  actuated  by  sound  energy  and  delivering 
electrical  energy.     (See  article  2,  above.) 

3.  Amplifiers  for  increasing  the  amplitude  of  this  electrical  energy.     (See  article  3 
above.) 

4.  Control  equipment  for  the  regulation  and  adjustment  of  this  electrical  energy.     (See 
articles  3  and  4  above.) 

5.  Wire  lines  to  carry  the  electrical  replica  of  the  original  sound  waves  from  the  studio 
to  the  radio  transmitter. 

6.  Radio  transmitter  for  converting  this  electrical  energy  into  radio-frequency  energy. 

7.  Antenna  system  for  radiating  the  radio-frequency  energy  into  space. 


16-26 


SOUND-EEPRODTJCTION  SYSTEMS 


A  schematic  diagram  of  a  typical  broadcast  transmitting  system,  sho-wing  the  general 
type  of  circuit  layout  employed,  is  given  in  Fig.  1.  A  single  studio  and  a  single  remote 
pick-up  point  are  represented,  each  such  point  requiring  a  duplicate  of  the  equipment 


FIG.  1.     Broadcasting  System  Layout 

shown,  up  to  and  including  the  relays  associated  -with  the  outgoing  lines  to  the  various 
networks. 

The  first  four  items  listed  above  as  parts  of  a  complete  broadcasting  system  have  been 
described  in  detail  in  the  opening  articles  of  this  section,  "Audio  Facilities  for  Sound 
Systems/*  The  remaining  items  listed  are  more  or  less  peculiar  to  broadcasting  systems 
and  are  covered  below. 


PROGRAM  LINES  16-27 


13.  PROGRAM  DISTRIBUTION  SYSTEMS 

NETWORK  SWITCHING  EQUIPMENT.  The  key  stations  of  a  network  of  broad- 
casting stations  must  provide  means  whereby  the  output  of  any  studio  may  be  distributed 
to  any  of  the  networks  or  combination  of  networks  radiating  out  from  the  city  in  which 
the  key  station  is  located.  Frequently  different  programs,  coming  from  different  origina- 
tion points,  are  simultaneously  sent  to  the  various  legs  o!  the  network  radiating  from  the 
key  station.  In  order  to  accomplish  these  operations  switching  means  are  used  whidb 
permit  the  connection  of  a  line  amplifier  across  the  program  bos  of  the  desired  program 
source. 

The  facilities  for  switching  are  usually  such  that  the  proper  studio  and  aetwork  line-op 
may  be  arranged  previous  to  "air  time"  but  without  actually  connecting  the  studio©  in- 
volved to  their  respective  networks  until  a  master  switch  is  operated.  Upon  the  proper 
cue,  or  at  the  proper  time,  operation  of  the  master  switch  connects  the  various  studios  iBt~ 
volved  to  the  right  networks.  The  actual  switching  is  seldom  accomplished  by  Biar};iml!y 
operating  the  switches  but  rather  through  the  medium  of  conveniently  located  relays 
which  are  remotely  controlled  from  the  operating  desk. 

BRIDGING  AMPLIFIER.  The  purpose  of  the  bridging  amplifier  is  to  isolate  the  out- 
going "radio"  lines  from  one  another  and  to  provide  a  means  of  connecting  any  number  of 
outgoing  lines  to  any  program  source,  at  will,  without  causing  any  unbalancing  or  imped- 
ance mismatch  of  the  equipment  line-up.  If  two  or  more  outgoing  radio  lines  feeding 
different  networks,  but  carrying  the  same  program,  were  to  be  connected  in  parallel  and 
thence  to  the  output  of  the  program  amplifier,  then,  should  any  noise,  ground,  or  otber 
fault  develop  on  one  line,  it  would  affect  the  operation  of  the  others.  By  placing  a  bridg- 
ing amplifier  (which  is,  of  course,  a  one-way  device)  in  each  line,  complete  isolation  is  ef- 
fected and  there  is  no  possibility  that  one  line  will  affect  others  being  fed  from  the  same 
studio. 

The  bridging  amplifier  also  permits  the  connection  of  any  reasonable  number  of  lines 
to  the  output  of  a  given  studio  amplifier  without  causing  an  impedance  mismatch  whicfe 
would  adversely  affect  the  operation  of  the  system.  To  accomplish  this  connection,  tiie 
output  of  the  program  amplifier  is  terminated  in  a  resistance  of  the  proper  size,  thereby 
presenting  a  practically  constant  load  for  the  amplifier.  The  input  impedance  of  the 
bridging  amplifier  is  then  made  very  high  and  is  "bridged"  across  the  desired  program  bus 
without  appreciably  affecting  the  load  impedance  being  presented  to  the  output  of  tfae 
program  amplifier. 

A  bridging  amplifier  is  associated  with  each  of  the  outgoing  lines  leading  to  a  local 
transmitter  or  to  a  network  of  stations. 

BIBLIOGRAPHY 

Chinn,  H.  A.,  CBS  Hollywood  Studios,  Proc.  I.R.M.,  July  1939,  p.  421. 
Rackey,  C.  A.,  Network  Broadcasting,  Elec.  Bng.,  January  1941 ,  p.  16. 

14.  PROGRAM  LINES 

Telephone  lines  are  employed  for  the  purpose  of  transmitting  a  program  from  one  studio 
or  station  to  another  station.  The  facilities  involved  may  be  divided  into  two  classes: 
(1)  local  lines  for  connecting  the  studio  remote  pick-up  points,  such  as  athletic  iieJds, 
theaters,  and  hotels,  and  also  those  lines  used  for  connecting  the  studios  to  the  kxml  trans- 
mitter; (2)  long  lines  interconnecting  a  network  of  transmitting  stations  throughout  the 
country. 

LOCAL  LINES.  When  the  stations  involved  are  In  the  same  city  the  line  connecting 
facilities  are  known  as  loops.  These  relatively  short  lines  may,  by  the  employment  of 
proper  terminal  equipment,  be  made  to  have  an  essentially  flat  transmission  vs.  frequency 
characteristic  over  the  entire  range  of  audio  frequencies  necessary  for  high-fidelity  broad- 
cast service  (see  Section  16,  article  18).  In  order  to  obtain  this  desirable  feature  the 
natural  attenuation  characteristics  of  the  lines,  which  for  tbe  most  part  are  cable  circmtfi, 
are  modified  at  the  receiving  terminals  by  means  of  an  attenuation  equaliser. 

LONG  LIKES.  If  the  stations  to  be  interconnected  are  in  different  eiti«s  the  connect- 
ing facilities  consist  of  special  telephone  lines  which  are  either  non-loaded  open  wire  or 
loaded  cable  circuits  (see  Section  17,  article  18).  Present  cable  facilities  are  loaded  a* 
intervals  slightly  in  excess  of  1/2  mile.  Amplifiers,  equalizers,  and  phase  correctors  are 


16-28  SOTJND-BEPBODTJCTION  SYSTEMS 

installed  at  approximately  50-mile  intervals  on  cable  circuits  and  about  125  miles  apart 
on  open  wire  facilities.  The  cable  circuits  have  automatic  regulators  installed  about 
every  150  miles  in  order  to  keep  the  loss  of  the  circuit  independent  of  the  temperature 
along  the  circuit. 

Attenuation  equalizers  are  employed  on  these  circuits  just  as  in  local  lines.  Velocity 
correctors  (see  Section  5,  article  10,  and  Section  17,  article  18)  are  also  necessary  to  com- 
pensate for  the  natural  characteristic  of  the  lines  which  results  in  an  unequal  time  delay 
in  the  transmission  of  the  various  component  waves  of  different  frequencies.  In  circuits 
less  than  500  miles  long  these  devices  would  not  be  necessary,  but  with  present  circuit 
requirements  of  2000  to  3000  miles  they  are  indispensable. 

Long-line  facilities  are  available  with  overall  transmission  vs.  frequency  characteristics 
that  are  essentially  uniform  over  the  entire  audio-frequency  range  necessary  for  high- 
quality  broadcasting. 

ATTENUATION  EQUALIZER.  (See  also  Section  17,  Article  18.)  An  attenuation 
equalizer  is  bridged  across  the  receiving  terminals  of  a  line  in  order  to  modify  the  natural 
characteristics  of  the  line  so  as  to  provide  a  circuit  having  an  essentially  uniform  trans- 
mission vs.  frequency  characteristic  over  the  range  of  frequencies  desired.  The  device  is 
connected  at  the  receiving  end  of  the  line  in  order  to  obtain  the  best  ratio  of  signal-to-noise 
and  interference  on  the  circuit. 

An  attenuation  equalizer  is  an  electrical  network  which  introduces  a  loss  at  each  fre- 
quency such  that  the  sum  of  the  line  and  equalizer  losses  is  the  same  for  all  frequencies  over 
the  useful  range. 

In  its  most  elementary  form  the  equalizer  consists  of  a  simple  resonant  circuit  in  series 
with  a  variable  resistance.  The  frequency  of  the  resonant  circuit  is  selected  so  that,  with 
the  proper  value  of  series  resistance,  the  overall  transmission  characteristic  of  the  circuit 
is  as  uniform  as  practical. 

BIBLIOGRAPHY 

Bode,  H.  W.,  Variable  Equalizers,  Bdl  Sys.  Tech.  J.,  April  1938,  p.  229. 

Clark,  A.  B.,  and  Green,  C.  W.,  Long  Distance  Cable  Circuit  for  Program  Transmissiom,  Bdl  Sys, 

Tech.  /.,  July  1930,  p.  567. 
Cowan,  F.  A.,  Telephone  Circuits  for  Program  Transmission,  Trans.  Am.  Inst.  of  Elec.  Eng.,  July 

1929,  p.  1045. 
Cowan,  McCurdy,  and  Lattimer,   Engineering  Requirements  for  Program  Transmission  Circuits. 

Bdl  Sys.  Tech.  J.,  April  1941,  p.  235. 

15.  BROADCASTING  TRANSMITTER  PLANT 

A  broadcasting  transmitting  plant  consists  of  audio  input  equipment,  modulator,  radio- 
frequency  generator  and  amplifier,  radio-frequency  transmission  line,  antenna  tuning 
equipment,  and  antenna  system. 

The  audio  equipment  associated  with  a  transmitter  plant  provides  facilities  for  such 
switching  operations  as  are  required,  microphone  and  turntable  equipment  for  local  pro- 
gram origination  in  an  emergency,  and  amplifiers  for  increasing  the  volume  level  of  the 
program  material  received  from  the  line  connecting  the  studios  to  the  transmitter.  After 
being  sufficiently  amplified  the  incoming  program  material  passes  to  the  modulator  tube 
which  modulates  the  radio-frequency  energy  generated  and  amplified  by  the  equipment 
supplied  for  that  purpose  (see  Section  7,  article  17;  also  Section  8,  article  4).  The  re- 
sultant modulated  radio-frequency  energy  may  either  be  further  amplified  or  sent  di- 
rectly to  the  antenna  tuning  equipment.  The  antenna  is  usually  located  a  relatively  short 
distance  from  the  building  housing  the  transmitter,  and  a  radio-frequency  transmission  line 
is  used  to  convey  the  energy  from  the  transmitter  to  the  antenna.  The  antenna  tuning 
equipment  is  usually  located  at  the  base  of  the  antenna  in  an  appropriate  protective  shelter. 

STANDARD  A-M  BROADCASTING  TRANSMITTING  ANTENNAS.  (See  Section  6, 
article  31.)  For  standard  broadcasting  (amplitude  modulation  in  the  540-1600  kc  band), 
the  vertical-radiator  antenna  is  generally  used.  The  use  of  an  antenna  having  an  elec- 
trical height  slightly  in  excess  of  0.5  wavelength,  and  operated  below  the  fundamental, 
results  in  the  largest  field  intensities  on  the  horizon  for  a  given  radiated  power.  At  the 
optimum  point  of  operation  the  electric  field  at  the  receiver  resulting  from  the  ground  wave 
radiated  by  the  antenna  may  be  as  much  as  40  per  cent  greater  than  that  obtained  from  a 
0.25-wave  antenna  radiating  the  same  power.  This  improvement  results  from  the  fact 
that  more  energy  is  radiated  along  the  ground,  where  it  is  desired,  and  less  up  in  the  air. 
It  does  not  follow  from  this,  however,  that  the  maximum  coverage  is  secured  by  an  antenna 
having  this  optimum  height. 


BROADCASTING  TRANSMITTER  PLANT  16-29 

For  the  low-powered  transmitter  where  the  primary  range  is  limited  by  the  field  in- 
tensity failing  below  the  prevailing  interference  level,  an  antenna  of  the  optimum  height 
would  probably  result  in  an  increased  service  area.  The  cost  of  such  a  radiator  in  compari- 
son with  the  cost  of  increasing  the  power  of  the  transmitter  sometimes  precludes  its  use, 
however. 

For  the  high-powered  transmitter  the  primary  range  is  generally  limited  to  that  distance 
where  "mushing"  results  from  the  admixture  of  the  ground  and  the  sky  wave,  at  this  point 
the  strength  of  the  waves  being  of  about  the  same  magnitude.  In  this  case  variations  in 
the  sky  wave  brought  about  by  varying  the  height  of  the  antenna  are  far  more  important 
in  determining  ^the  primary  range  of  the  station  than  attendant  variations  in  the  ground 
wave.  The  height  that  is  the  best  operating  point  for  the  greatest  ground-wave  intensity 
is  not  always  the  best  height  from  the  viewpoint  of  pushing  out  the  incipient  fading  dis- 
tance by  the  reduction  of  the  sky  wave.  Hence,  the  best  operating  condition  for  maxi- 
mum primary  coverage  is  not  necessarily  that  height  which  results  in  the  maximum  ground 
wave.  The  best  height  for  a  given  antenna  depends  upon  the  attenuation  of  the  ground 
wave,  which  in  turn  depends  upon  the  effective  conductivity  of  the  soil,  its  dielectric  con- 
stant, and  the  frequency  of  operation.  In  any  event  the  optimum  electrical  height  is 
likely  to  be  between  0.5  and  0.6  wavelength. 

The  economical  advantage  of  an  antenna  of  this  height  depends  upon  the  transmitted 
power.  The  initial  investment  and  the  cost  of  operation  of  the  transmitting  plant  in- 
crease with  the  power,  whereas  the  cost  of  the  radiating  structure  remains  practically  con- 
stant. At  the  higher  powers  this  type  of  radiator  represents  a  good  balance  between  the 
two  investments. 

DIRECTIONAL  ANTENNAS  FOR  STANDARD  (A-M)  BROADCASTING.  The  ap- 
plication of  antenna  systems  having  definite  directional  properties  to  broadcasting  pur- 
poses has  been  undertaken  in  a  number  of  instances.  Among  the  circumstances  which 
have  led  to  the  installation  of  a  directional  system  are:  the  need  for  suppressing  radiation 
in  a  particular  direction  or  directions  in  order  to  prevent  interference  with  a  distant  station 
or  stations  operating  on  the  same  channel;  the  desire  to  suppress  radiation  in  a  given  di- 
rection where  no  audience  exists  and  to  reinforce  transmission  towards  the  populated  area, 
as  for  instance  in  a  station  located  on  a  seacoast  or  to  one  side  of  a  town  which  constituted 
its  principal  audience. 

A  suitable  number  of  vertical  antenna  elements  properly  phased  and  spaced  are  usually 
employed  in  order  to  obtain  the  desired  directional  characteristic.  By  the  proper  com- 
bination of  these  antenna  elements  and  their  proper  phasing  almost  any  desired  direc- 
tional pattern  may  be  obtained  (see  Section  6,  article  29).  Either  vertical- wire  antennas 
or  towers  insulated  at  their  base  are  used  for  the  antenna  elements. 

F-M  BROADCASTING  TRANSMITTER  ANTENNAS.  For  f-m  broadcasting  (fre- 
quency modulation  in  the  88-  to  lOS-Mc  band),  a  horizontally  polarized  antenna  system 
is  employed.  In  general  the  antenna  is  non-directional  in  a  horizontal  plane.  However, 
since  most  receiving  sites  are  located  within  a  few  degrees  of  the  horizon,  it  is  advantageous 
to  utilize  an  antenna  system  which  directs  the  radiation  towards  the  horizon.  The  gain 
realized  by  this  practice  permits  the  use  of  lower  actual  transmitter  power  for  a  given 
"effective"  radiated  power  (effective  radiated  power  is  the  actual  power  multiplied  by  the 
power  gain  of  the  antenna  in  the  direction  of  the  horizon).  In  practice,  the  cost  of  trans- 
mitters of  various  power  levels  must  be  balanced  against  the  cost  of  directional  antennas 
of  various  gains  in  determining  the  optimum  combination. 

The  f-m  broadcasting  antenna  must  be  located  at  a  point  of  high  elevation  in  order  to 
reduce  to  a  minimum  the  shadow  effect  on  propagation  of  hills  and  buildings.  To  provide 
the  best  service  to  an  area,  a  high  antenna  is  usually  preferable  to  a  lower  one  with  in- 
creased transmitter  power. 

STANDARD  (A-M)  BROADCAST  STATION  TRANSMITTER  SITES.  The  selection 
of  a  good  site  for  a  standard  (a-m)  broadcast  transmitter  is  a  very  complex  problem  which 
involves  many  considerations.  The  site  should  be  selected  with  the  view  to  providing: 

1.  Satisfactory  coverage  of  the  area  comprising  the  population  it  is  desired  to  serve. 
Usually  this  consideration  will  fix  the  maximum  distance  from  the  center  of  the  city  that 
the  transmitter  can  be  located. 

2.  Maximum  coverage  of  adjacent  populated  areas  consistent  with  fulfilling  the  above 
requirement. 

3.  Minimum  population  in  the  area  immediately  adjacent  to  the  transmitter  where  the 
signal  is  likely  to  be  so  strong  that  special  precautions  may  have  to  be  taken  to  insure  good 
reception  from  other  stations. 

4.  Good  soil  conditions  at  the  transmitter  site.     The  conductivity  of  the  soil  within 
several  wavelengths  of  the  antenna  has  considerable  bearing  upon  the  efficiency  of  the 
antenna  and  the  nature  of  its  radiation  characteristics. 


16-30  SOUND-EEPKODUCTION  SYSTEMS 

5.  Good  power  and  program  circuit  facilities.     If  possible,  two  sources  of  power  coming 
from  different  directions  should  be  obtained.     In  order  to  obtain  better  regulation,  it  is 
often  advisable  to  obtain  power  from  a  high-voltage  line  and  have  a  local  substation  in- 
stalled.    Because  of  their  relative  immunity  from  storms,  telephone  lines  in  cable  should 
be  obtained. 

6.  Low  cost  of  land.     The  size  of  the  plot  necessary  will  depend  upon  the  size  of  the 
ground  system,  the  spacing  of  the  towers,  and  the  distance  between  the  anchors  for  the 
guys. 

7.  Good  publicity  value  and  accessibility.     These  are  good  assets  for  a  station  but  may 
be  overemphasized.     Of  course,  roads  leading  to  the  transmitter  should  be  usable  in  any 
kind  of  weather. 

8.  Immunity  from  floods,  storms,  sleet,  etc.,  whenever  possible,  and  ground  suitable 
for  good  tower  foundations.     In  some  instances  severe  storms  are  localized  in  certain  areas 
that  can  be  avoided-     Severe  storms  may  cripple  power  and  telephone  facilities. 

9.  Proper  location  with  respect  to  airports  and  airways. 

10.  Proper  location  with  respect  to  large  metal  obstructions,  buildings,  etc. 

F-M  BROADCAST  STATION"  TRANSMITTER  SITES.  The  selection  of  a  site  for  a 
f-m  broadcasting  station  entails  considerations  somewhat  different  than  those  for  standard 
broadcast  stations.  Many  of  the  differences  stem  from  the  quasi-optical  nature  of  the 
very-high-frequency-wave  propagation.  The  transmitter  site  should  be  chosen  with  these 
factors  in  mind: 

1.  The  location  should  be  as  near  the  center  of  the  proposed  service  area  as  possible 
consistent  with  the  availability  of  a  site  with  sufficient  elevation  to  provide  service  through- 
out the  area. 

2.  The  location  should  provide  Hne-of-sight  over  the  principal  city  or  cities  to  be  served. 
No  major  obstructions  should  be  in  the  path. 

3.  The  site  should  be  so  situated  that  the  field  intensity  in  the  urban  area  is  sufficiently 
great  to  provide  satisfactory  service  in  spite  of  the  generally  higher  electrical  interference 
in  such  areas. 

4.  Good  power  and  program  circuit  facilities  are  required. 

5.  If  the  cite  is  a  high  building,  consideration  must  be  given  to  the  problems  of  installing 
the  antenna  and  the  transmitter. 

6.  Cognizance  must  be  taken  of  the  possible  hazard  of  the  antenna  to  aviation. 

BIBLIOGRAPHY 

Brown,  G.  H.f  Directional  Antennas,  Proc.  I.R.E.,  January  1937,  p.  78. 

Chamberlain,  A.  B.,  and  Lodge,  W.  B.,  The  Broadcast  Antenna,  Proc.  I.R.E.,  January  1936,  p.  11. 

Lodge,  W.  B.,  The  Selection  of  a  Radio-broadcast  Transmitter  Site,  Proc.  I.R.E.,  October  1939,  p.  621. 

16.  BROADCAST  FREQUENCY  ALLOCATION 

The  frequency  spectrum  now  known  to  the  radio  art  extends  over  a  wide  range  and  in- 
cludes frequencies  having  widely  different  characteristics.  This  spectrum  is  occupied  not 
only  by  broadcasting  services  but  also  by  other  kinds  of  radio  services  such,  as  communica- 
tion with  ships  and  aircraft,  police  services,  and  amateur,  experimental,  transoceanic,  and 
transcontinental  point-to-point  communication,  both  telegraph  and  telephone.  Explor- 
atory work  is  still  going  on  in  the  higher  frequencies  at  the  upper  end  of  the  spectrum  and 
is  directed  in  part  to  determining  their  usefulness  for  broadcasting  purposes. 

The  nations  of  the  world  have  agreed  to  devote  certain  portions  of  the  radio-frequency 
spectrum  to  broadcasting  purposes.  The  standard  a-m  (amplitude  modulation)  broad- 
cast band  extends  from  540  to  1600  kc  per  see  and  is  used  generally  throughout  the  world. 
A  band  extending  from  160  to  265  kc  is  used  in  Europe  but  not  in  t.M«  country  for  a-m 
broadcasting.  Several  narrow  bands  in  the  high-frequency  spectrum  (above  6000  kc) 
are  also  in  use  for  long-distance  a-m  broadcasting  services.  Finally,  a  band  extending 
from  88  to  108  Me  is  used  in  this  country  for  f-m  (frequency  modulation)  broadcasting. 
The  wave  propagation  characteristics  of  transmissions  made  in  these  various  bands  dif- 
fer radically  (see  Section  10,  article  24). 

STANDARD  BROADCASTING.  The  term  "standard  broadcasting"  is  applied  to  a-m 
stations  operating  in  the  band  of  frequencies  from  540  to  1600  ke.  Each  station  is  assigned 
a  particular  carrier  frequency.  On  the  North  American  continent  the  assignable  fre- 
qraeneies  extend  throughout  the  range  in  10-kc  intervals.  Thus  the  assignments  are  540, 
550, 560,  etc.,  up  to  1600  kc,  making  a  total  of  107  distinct  channels. 

There  are  three  classes  of  standard  broadcast  channels:  clear,  regional,  and  local. 


BIBLIOGRAPHY 

^A  clear  channel  is  one  on  which  the  dominant  station  or  stations  render  service  over 
wide  areas  and  which  are  cleared  of  objections!  interference  within  their  ground-wave 
service  areas  and  over  all  or  a  substantial  portion  of  their  sky-wave  service  areas. 

A  regional  channel  is  one  on  which  several  stations  may  operate  with  powers  not  in 
excess  of  5  kw.  The  ground-wave  service  area  of  a  station  operating  on  any  such  channel 
may  be  limited,  as  a  consequence  of  interference,  to  a  given  field-intensity  contour. 

A  local  channel  is  one  on  which  several  stations  may  operate  with  powers  not  in  excess 
of  250  watts.  The  ground-wave  service  area  of  a  station  operating  on  any  such  channel 
may  be  limited,  as  a  consequence  of  interference,  to  a  gives  field-intensity  contour. 

By  assigning  adjacent  channels  in  widely  separated  areas  of  the  country  potential  in- 
terference is  minimized.  In  any  one  area  it  is  common  practice  to  separate  the  channels 
by  approximately  30  kc.  This  leaves  sufficient  frequency  separation  to  enable  receiving 
sets  to  select  one  channel  to  the  exclusion  of  all  others  in  that  area, 

HIGH-FREQUENCY  BROADCASTIHG.  By  international  agreement  high-frequency 
bands  have  been  allocated  for  broadcasting  services  in  the  vicinity  of  6,  9,  11,  15,  17,  and 
21  Me.  Transmissions  in  these  bands  are  utilized  for  a  purpose  and  in  a  manner  entirely 
different  from  those  in  the  a-m  or  f-m  broadcast  bands.  High-frequency  transmissions 
are  primarily  intended  for  long-distance  broadcasts  to  distant  colonial  possessions,  iso- 
lated territories,  and  overseas  broadcasting.  This  type  of  service  depends  entirely  upon 
the  sky  wave  for  reception  as  contrasted  to  regular  broadcast  transmissions  which  utilize 
the  ground  wave  for  primary  coverage  (see  below).  High-frequency  transmission  to  dis- 
tant points  is  not  very  satisfactory  when  reception  is  obtained  with  the  relatively  simple 
equipment  available  for  the  broadcast  listener.  Magnetic  disturbances  and  atmospheric 
conditions  seriously  affect  high-frequency  transmissions  and  cause  amplitude  and  seietrtive 
fading  and  associated  deterioration  of  tonal  quality. 

F-M  (FREQUENCY  MODTTLATIOH)  BROABCASTEfG.  Frequendes  above  30,000 
kc  are  referred  to  as  very  high  frequencies.  These  wave®  are  sometimes  kaown  as  quasi- 
optical  waves  because  their  transmission  characterisfcies  resemble,  in  many  respects,  those 
of  visible  light  waves  (see  Section  10,  article  20).  As  a  consequence  tfee  service  range  of 
a  very-high-frequency  broadcasting  station,  evess  if  located  on  a  high  point  so  that  the 
waves  travel  to  the  receiving  station  with  a  minimum  of  obstacles  in  their  path,  is  limited 
to  several  tens  of  miles. 

As  compared  with  standard  a-m  broadcasting  frequencies,  very  high  frequencies  present 
several  advantages.  Interference  caused  by  natural  atmospheric  disturbances  (static) 
is  essentially  non-existent,  and  therefore  reception  is  markedly  less  dependent  on  seasonal 
influences.  The  service  range  of  the  station  is  more  clearly  defined  and  independent  of 
any  normal  Heaviside  layer  conditions.  The  area  over  which  a  very4dgji-fr«quency  sta- 
tion creates  interference  with  other  stations  on  the  same  or  adjacent  frequencies  is  not 
so  great,  compared  to  the  useful  service  area,  as  in  standard  broadcasting  frequencies. 
A  substantial  advantage  exists  in  this  respect  that  is  of  real  assistance  in  v^ry-high-fre- 
quency  allocation.  The  dimensions  of  the  receiving  antenna  can  be  small,  aad  direc- 
tional transmission  and  reception  are  relatively  easy. 

The  disadvantages  of  very-high-frequency  waves  are  inherent  in-  their  very  nature.  The 
high  absorption  during  propagation  limits  tfee  service  range  so-  tbat  the  covering  of  a  large 
geographical  area  by  this  means  oa  an  economical  basis  presents  a  problem.  Because  of 
the  quasi-optical  character  of  the  very-high-frequency  waves  there  may  be  propagation 
shadows  and  areas  of  relatively  poor  reception,  particularly  near  hilly  terrain  or  high 
buildings. 

In  this  country  the  band  of  frequencies  from  S8  to  108  Me  has  been  assigned  for  very- 
high-frequency  broadcasting.  The  assignable  frequencies  extend  throughout  the  range 
in  200-kc  intervals.  Thus  the  assignments  are  88. 1,  88.3,  88.5,  etc,,  up  to  107.9  Me, 
making  a  total  of  100  distinct  channels.  Frequency  modulation,  with  a  carrier  swing  of 
±75  kc,  is  used.  The  term  f-m  broadcasting  is  applied  to  this  class  of  service. 

Currently  in  this  country  there  are  two  classes  of  f-m  stations.  Those  designated  as 
Class  A  are  designed  to  render  service  primarily  to  a  community  or  U>  a  city  or  town  other 
than  the  principal  city  of  the  area  and  the  surrounding  rural  area.  Class  B  stations  are 
designed  to  render  service  primarily  to  metropolitan  districts  or  principal  cities  and  sur- 
rounding rural  area,  or  to  rural  areas  removed  from  large  ©enters  of  populatkni. 

BIBLIOGRAPHY 

Federal  Communications  Commission,  Standard*  of  Good  Engineering  -Praxes  coaming  Standard 
and  FM  Broadcast  Stations.    Federal  Communications  Communion  Rules  and  Regulations. 


16-32  SOUND-REPRODUCTION  SYSTEMS 


17.  BROADCASTING  STATION  SERVICE 

STANDARD  BROADCAST  COVERAGE.  In  considering  the  probable  service  area 
of  a  standard  broadcast  station  it  is  necessary  to  take  into  account  the  effects  of  both  the 
ground  and  sky  waves  which  are  radiated  by  the  transmitting  antenna. 

The  ground  wave  (or  direct  ray)  which  travels  directly  over  the  surface  of  the  earth 
from  the  transmitter  to  the  receiver  is  unaffected  in  its  propagation  by  meteorological  or 
seasonal  conditions  and  is  of  the  same  intensity  during  both  the  day  and  the  night.  The 
sky  wave  (or  indirect  ray)  which  traverses  the  Heaviside  layer  is  subject  to  a  great  deal  of 
variation  in  strength  and  character  before  reaching  the  receiving  point  (see  Section  10, 
article  24). 

The  radiated  energy  which  follows  close  to  the  earth,  called  the  ground  wave,  is  char- 
acterized by:  (a)  high  field  intensities  near  the  transmitter;  (6)  attenuation  to  low  values 
within  a  few  tens  of  miles,  depending  upon  the  character  of  the  ground,  the  power  and 
frequency  of  the  transmitter  signal,  and  the  type  of  transmitting  antenna;  (c)  relatively 
steady  values. 

The  energy  which  is  reflected  back  from  the  ionosphere  (chiefly  evident  after  sunset), 
called  the  sky  wave,  is  characterized  by:  (a)  considerable  field  intensity  at  distances  of 
hundreds  of  miles  from  the  transmitter;  (&)  wide  variation  in  field  intensity  from  moment 
to  moment,  from  night  to  night,  and  from  year  to  year;  (c)  considerable  variation  with 
latitude  of  the  transmission  path  and  the  earth  characteristics  in  the  vicinity  of  the  trans- 
mitter, and  some  variation  with  frequency. 

Because  of  its  steady  nature  and  the  strong  signals  obtainable  in  areas  near  the  trans- 
mitter, the  preferable  service  of  any  station  is  that  obtained  from  ground  waves.  The 
extent  of  the  areas  of  ground-wave  service  is  determined  not  only  by  the  transmitter 
power,  by  the  frequency  and  type  of  antenna,  and  by  the  ground  conductivity  in  the  area, 
but  also  by  the  interference  to  the  desired  signal  caused  by  atmospheric  noise,  man-made 
noise,  other  stations  on  the  same  or  adjacent  channels,  or,  under  certain  conditions  at 
night,  by  the  fading  and  distortion  caused  by  a  mixture  of  ground  wave  and  sky  wave. 
Since  the  intensity  of  these  limiting  factors  varies  widely  from  moment  to  moment  and 
from  night  to  night,  the  area  of  satisfactory  service  also  varies. 

Since  the  strong  steady  ground-wave  service  is  in  general  available  only  within  a  rel- 
atively short  distance  of  the  transmitter,  a  considerable  part  of  the  country  lies  outside 
such  areas.  This  is  particularly  true  at  night,  when  strong  sky-wave  signals  from  distant 
stations  on  the  same  channel  cause  considerable  interference  in  many  instances  and  thus 
reduce  the  effective  service  area  from  its  daytime  value.  At  night  in  these  areas  use  can 
be  made  of  any  interference-free  sky-wave  signals  for  service,  but,  because  of  its  wide 
variation  in  intensity  and  occasional  periods  of  signal  distortion,  such  service  is  considered 
in  general  less  desirable  than  ground-wave  service.  The  extent  of  the  area  of  satisfactory 
sky-wave  service  depends  upon  the  interference  to  the  desired  signal  caused  by  atmospheric 
and  man-made  noise,  other  stations  on  the  same  and  adjacent  channels,  and  under  certain 
conditions  by  the  fading  and  distortion  caused  by  a  mixture  of  the  station's  own  ground- 
and  sky-wave  signals.  In  view  of  the  wide  variation  in  sky-wave-signal  level  from  night  to 
night,  the  area  in  which  satisfactory  listening  can  be  had  on  any  one  night  will  vary  greatly. 

The  zone  in  which  fading  is  first  encountered  is  at  that  distance  where  the  sky  wave 
becomes  of  such  intensity  as  to  interfere  with  the  ground  wave.  The  primary  range  of 
the  station  may  be  extended  either  by  increasing  the  strength  of  the  ground  wave  or  by 
decreasing  the  strength  of  the  sky  wave  (see  Section  10,  article  23).  If  the  receiver  is 
located  within  the  incipient  fading  distance  an  increase  in  transmitting  power  above  the 
noise  level  improves  reception  and  increases  the  primary  range.  In  the  fading  zone, 
however,  an  increase  in  power  beyond  that  required  to  produce  an  average  field  intensity 
sufficient  to  override  noise  produces  no  further  increase  in  service  area.  This  is  because  the 
strength  of  the  ground  wave  and  that  of  the  sky  wave  are  being  increased  simultaneously 
and  thus  the  relationship  between  them  is  maintained  constant.  It  is  therefore  evident 
that  with  the  low-powered  transmitter  the  primary  range  is  limited  by  the  field  intensity 
falling  below  the  prevailing  noise  level  at  the  receiving  point.  With  the  high-power  trans- 
mitter the  primary  range  is  more  likely  to  be  limited  by  fading  and  attendant  objection- 
able phenomena  since  the  ground  wave  will  usually  be  strong  enough  to  override  noise  out 
to  and  beyond  the  point  where  fading  begins.  For  detailed  calculations  of  broadcast 
station  range  see  Section  10,  article  24. 

F-M  BROADCASTING  COVERAGE.  Although  some  service  may  be  provided  by 
troposphenc  waves,  the  service  area  of  a  f-m  broadcasting  station  is  considered  to  be  only 
that  served  by  the  ground  wave.  The  extent  of  the  service  area  is  determined  by  the  point 
at  which  the  ground  wave  is  no  longer  of  sufficient  intensity  to  provide  satisfactory  re- 


FIDELITY  EEQUmEMENTS  OF  BROADCAST  SYSTEM      16-33 

ception.  The  field  intensity  necessary  for  service  in  city,  business,  or  factory  areas  is 
generally  considered  to  be  1000  microvolts  per  meter.  In  rural  areas,  on  the  other  hand, 
50  microvolts  per  meter  is  generally  believed  to  be  sufficient  for  good  reception.  These 
figures  are  based  upon  the  usual  noise  levels  encountered  and  upon  the  absence  of  inter- 
ference from  other  stations. 

The  ground-wave-signal  range  of  a  f-m  broadcasting  station  is  a  function  of  the  heights 
of  the  transmitting  and  receiving  antennas,  the  gain  of  the  antennas,  the  transmitter 
power,  the  frequency,  the  ground  conductivity,  and  the  dielectric  constant.  The  service 
area  of  a  f-m  station,  just  like  that  of  standard  a-m  broadcasting  stations,  may  be  ac- 
curately calculated  by  known  methods  (see  Section  10,  article  20).  A  detailed  study  of 
the  service  areas  possible  with  f-m  broadcast  ing  stations  develops  these  facts: 

1.  The  service  area  of  approximately  half  of  all  United  States  low-power  (100-  and  250- 
watt)  standard  a-m  broadcast  stations  could  be  increased  by  going  to  frequency  modula- 
tion. 

2.  Standard  broadcast  stations  hi  areas  of  poor  soil  conductivity  would  benefit  by  a 
change  to  frequency  modulation. 

3.  Standard  broadcast  stations  having  frequency  assignments  in  the  high-freqiiency 
end  of  the  band  would  gain  by  a  change  to  frequency  modulation. 

INTERFERENCE  TO  BROADCAST  SERVICE.  The  strength  of  the  electric  field 
produced  at  the  receiving  location  depends  on  many  factors  such  as  the  power  of  the 
transmitting  station,  the  nature  and  efficiency  of  the  antenna  system,  the  distance  in- 
volved, the  nature  of  the  intervening  terrain,  and  in  some  cases  the  time  of  the  day  and 
the  season  of  the  year.  At  a  particular  receiving  location,  in  addition  to  the  electric  field 
strength  produced  by  the  desired  broadcast  station,  other  electric  fields  will  exist  which 
may  hamper  or  prevent  reception,  It  is  not  the  absolute  electric  field  strength  produced 
by  the  desired  broadcast  station  which  determines  whether  reception  will  be  satisfactory; 
it  is  the  ratio  of  the  desired  field  strength  to  the  predominating  interfering  fields,  coupled 
with  the  ability  of  the  receiving  set  to  diseriininate  against  those  interfering  fields,  which 
determines  the  success  of  the  reception. 

Interfering  fields  may  arise  from  atmospheric  disturbances  (static),  from  industrial 
electrical  interference,  and  from  stations  operating  on  the  same  or  different  channels. 

The  intensity  of  the  atmospheric  noise  is  not  constant  throughout  the  radio-frequency 
spectrum.  At  night  it  varies  inversely  with  the  frequency;  daytime  atmospheric  noise 
varies  approximately  inversely  as  the  square  of  the  frequency.  The  magnitude  of  the 
noise  depends  upon  the  geographical  location  of  the  receiving  point,  the  season  of  the  year, 
and  the  conditions  existing  at  the  receiver. 

Industrial  electrical  interference  produced  by  the  operation  of  non-radio  electrical  de- 
vices is,  on  the  average,  inversely  proportional  to  the  radio  frequency.  There  are  also 
present  within  the  receiving  apparatus,  itself,  sources  of  noise  that  require  consideration: 
resistor  noise,  tube  noise,  contact  noise,  and  noise  associated  with  the  tube  power  supply. 
In  general,  this  receiving-set  noise  is  independent  of  the  radio  frequency  to  which  the  re- 
ceiver is  tuned, 

CONTINUITY  OF  SERVICE.  One  of  the  prime  prerequisites  for  the  successful 
operation  of  a  broadcasting  station  is  the  absolute  continuity  of  service  throughout  the 
broadcast  day.  The  necessity  for  such  operation  arises  from  the  keen  competition  among 
the  many  stations  in  this  country  and  the  psychological  reaction  of  the  average  listener 
to  an  interruption  in  the  program  service. 

This  requirement  imposes  a  severe  responsibility  upon  the  equipment  and  on  the  main- 
tenance crew  of  a  broadcasting  station,  inasmuch  as  the  stations  at  the  broadcasting 
centers  and  those  associated  with  the  major  chains  operate  from  16  to  IS  hours  con- 
tinuously every  day  of  the  year.  In  planning  a  broadcasting  station  many  precautionary 
measures  must  be  taken  and  suitable  devices  must  be  provided  to  permit  the  instant 
isolation  and  replacement  of  any  equipment  that  becomes  defective  during  the  course  of 
operation. 

18.  FIDELITY  REQUIREMENTS  OF  BROADCAST  SYSTEM 

A  high-quality  broadcasting  system  is  one  which  acoustically  transports  the  listener  in 
fancy  from  his  loudspeaker  to  the  studio  or  auditorium.  It  must  be  free  from  frequency, 
non-linear,  and  velocity  distortion.  It  must  not  introduce  extraneous  sounds  of  annoying 
nature  or  distracting  magnitudes. 

TONAL  RANGE.  Complete  freedom  from  frequency  distortion  implies  that  the  system 
should  be  uniformly  responsive  over  the  entire  range  of  audible  frequencies.  The  audible 
range  of  the  ear  depends  upon  a  great  many  factors  (see  Section  12,  article  2),  the  ex- 


16-34  SOTJND-REPBODUCTION  SYSTEMS 

treme  limits  being  in  the  neighborhood  of  16  to  16,000  cycles  per  second — some  10  octaves. 
In  practice,  few  persons  can  hear  this  extreme  range  at  the  listening  levels  normally  used 
in  the  home  (65-75  db  above  the  acoustical  reference  level  of  10~16  watt  per  cm2).  Fur- 
thermore, studies  seem  to  indicate  that  few  care  to  hear  this  extreme  range  when  listening 
to  broadcast  music. 

Several  kinds  of  investigations  have  been  undertaken  to  obtain  data  that  would  be  of 
assistance  in  determining  the  optimum  tonal  range  of  a  practical  system.  One  series  of 
tests  was  based  on  the  acuity  of  hearing  of  the  listeners.  These  experiments  were  con- 
cerned only  with  the  physical  ability  to  hear  differences  in  band  width;  they  disregarded 
the  question  of  the  enjoyment  or  esthetic  appreciation  of  wider  bands.  The  types  of  pro- 
gram material  included  a  dance  orchestra,  two  symphony  orchestras,  male  speech,  and  a 
dramatic  sketch.  The  observers  were  engineers  who  had  had  extensive  experience  in 
tests  of  program  quality  and  were  considerably  more  critical  than  the  average  radio  listener. 
It  was  found  that  a  change  of  band  width  from  15  to  8  kc  had  to  be  made  to  be  as  readily 
detected  as  a  change  from  8  to  5  kc.  These  changes,  for  speech,  are  just  sufficient  to  have 
an  equal  chance  of  being  detected  by  listeners  having  experience  in  such  tests. 

Changes  in  band  width  were  found  to  be  about  twice  as  readily  detected  with  music 
as  with  speech.  Thus,  for  music,  the  changes  that  were  just  discernible  half  of  the  time 
were  found  to  be  15  to  11  kc,  11  to  8  kc,  8  to  6.5  kc,  and  6.5  to  5  kc. 

In  another  kind  of  test  the  tonal  range  preferences  of  a  cross-section  of  radio  broadcast 
listeners  were  studied.  As  contrasted  to  the  studies  that  have  been  made  to  determine 
the  ability  to  distinguish  between  different  band  widths,  this  undertaking  ascertained  the 
tonal  range  that  the  average  listener  considered  most  pleasant,  that  is,  the  method  of 
reproduction  the  listener  would  select  in  his  home  when  listening  for  enjoyment.  Classic 
and  popular  music,  male  and  female  speech,  piano,  and  mixed  voices  with  sound  effects 
were  employed.  Every  possible  precaution  was  taken  during  the  tests  to  remove  any 
possibility  of  factors  other  than  tonal  range  from  influencing  the  listeners.  A  noise-freer 
essentially  distortionless  system  was  used,  and  the  reproduction  level  was  adjusted  in  ac- 
cordance with  the  listener's  desires. 

In  these  tests  the  cut-off  of  both  the  low  and  the  high  frequencies  was  gradual,  in  keep- 
ing with  the  type  found  in  actual  radio  receivers.  It  was  found  that  listeners  preferred  a 
tonal  range  whose  upper  frequency  limit  was  down  about  3  db  at  5000  cycles,  about  20  db 
at  8000  cycles,  and  about  30  db  at  10,000  cycles  with  respect  to  the  mid-range  frequencies. 
(The  experiments  did  not  test  the  preference  for  different  rates  of  cut-off  but,  rather,  for 
different  tonal  ranges  all  having  cut-offs  of  about  the  same  rate  as  the  one  mentioned.) 
Most  listeners  preferred  a  limited  tonal  range  to  a  wider  one  even  when  told  that  one  con- 
dition was  representive  of  "low  fidelity"  and  the  other  of  "high  fidelity.'* 

In  practice,  broadcast  transmitting  systems  are  designed  to  provide  uniform  trans- 
mission over  a  wide  range  of  frequencies.  A-m  broadcast  transmitters,  for  example,  are 
capable  of  covering  the  audio  spectrum  from  50  to  at  least  10,000  cycles,  with  negligible 
variations.  F-m  broadcast  transmitters  cover  a  still  wider  band,  extending  to  at  least 
15,000  cps.  On  the  other  hand,  except  for  a  few  isolated  Instances,  commercially  avail- 
able receiving  sets  are  not  capable  of  faithfully  reproducing  anywhere  near  this  range  of 
frequencies. 

Intercity  network  wire  facilities  having  a  very  uniform  frequency  characteristic,  par- 
ticularly at  the  higher  frequencies,  can  be  secured,  but  their  general  use  is  a  matter  of 
economic  consideration.  For  all  practical  purposes,  the  overall  frequency-response  char- 
acteristics of  a  complete  broadcasting  system  is  limited  by  the  wire  line  characteristics. 

The  rate  at  which  high-fidelity  receiving  equipment  is  put  into  service  will,  to  a  great 
extent,  influence  the  employment  of  better  wire  line  facilities  between  studios  and  radio 
stations. 

DYNAMIC  VOLUME  RAKGE.  The  dynamic  volume  range  of  a  sound  source  of 
varying  intensity  is  the  ratio  of  the  loudest  sound  produced  to  the  minfmnm  sound  that  is 
distinguishable.  In  broadcasting  and  sound  recording,  the  loudest  sound  intensities  are 
usually  experienced  with  symphonic  orchestras  or  special  sounds  such  as  explosions,  gun- 
fire, and  factory  noises  (see  Section  12) .  The  mininium.  audible  sound  intensity  is  a  func- 
tion of  the  residual  noise  level. 

i  As  noted  above  (article  1),  the  maximum  sound  intensities  encountered  in  studios  is 
about  95  db  for  music.  Furthermore,  it  was  stated  that  room  noise  levels  of  25  db  are 
generally  considered  satisfactory.  Thus  it  is  evident  that  the  ma-yin-mm  dynamic  range 
likely  to  be  encountered  in  original  performances  is  about  70  db  (excluding  special  sounds 
which  may  reach  any  intensity) .  This  is  a  somewhat  wider  range  (about  10  db)  than  can 
bfe  accommodated  by  most  complete  sound-reproducing  systems.  It  is  considerably  wider 
than  the  range  that  listeners  prefer. 

Very  few  listening  environments  are  capable  of  making  full  use  of  even  a  60-db  dynamic 
range.  In  the  home,  for  example,  the  average  listening  level  is  between  65  and  70  db  above 


POLICE   RADIO  16-35 

the  acoustical  reference.  The  residual  noise  level,  on  the  other  hand,  is  43  dfa  in  the 
average  residence,  and  in  only  1  per  cent  of  the  homes  is  it  as  Sow  as  30  db.  Thus,  even  in 
the  quietest  suburban  homes  the  noise  level  is  about  40  db  below  the  average  listening  level 
and  in  the  average  home  about  30  db  down.  However,  the  source  of  the  noise  is  not  likely 
to  be  in  the  same  direction  from  the  listener  as  the  radio  receiver.  Consequently,  the 
benefits  of  binaural  hearing  (article  1,  above)  will  assist  the  listener  in  partially  disregard- 
ing room  noise.  Nevertheless,  at  the  average  listening  level,  a  60-db  dynamic  range  can- 
not be  fully  exploited  by  the  listener  even  in  the  quietest  homes. 

As  a  corollary  to  the  question  of  dynamic  range,  it  has  been  found,  by  studies  in  which  a 
cross-section  of  broadcast  listeners  participated,  that  listeners  prefer*  to  hear  music  and 
speech  at  about  the  same  peak  levels  (as  read  by  a  standard  volume  indicator,  see  article 
4,  above).  It  was  also  found  that  the  limit  of  the  range  of  peo&  volume  levels  tolerated 
by  the  largest  number  of  listeners  is  approximately  8  db  (4  db  above  and  below  the  average 
volume  level  of  the  program).  Even  within  this  8-db  range  it  appears  that  changes  in 
volume  level  are  less  annoying  when  made  gradually.  The  S-db  limit  refers  to  the  range 
of  peak  or  maximum  volume  levels,  not  to  the  range  of  minimum  and  maximum  sound  in- 
tensities or  "dynamic  range."  (It  is  important  that  this  preferred  range  in  peak  levels 
not  to  be  confused  with  dynamic  range,  which  was  discussed  in  the  opening  paragraphs  of 
this  section.) 

It  was  also  found  that,  regardless  of  the  absolute  sound  intensity  at  which  the  listener 
operates  his  radio,  he  still  prefers  an  even  peak  intensity  level.  This  is  true  whether  he 
is  listening  to  variety,  drama,  narrative,  or  musical  programs,  The  peak  intensities  of 
the  main  program  material  (but  not  necessarily  background  effects)  must  not  fall  more 
than  8  db  below  the  maximum  peak  level;  otherwise  the  conditions  for  easy  listening  are 
violated. 

BIBLIOORAPHY 

Chinn,  H.  A.,  and  P.  Eisenberg,  Tonal-range  and  Sound-intensity  Preferences  of  Broadcast  Lfeteners, 

Proc.  I.R.E.,  September  1945,  p.  571. 

Chirm,  H.  A.,  and  P.  Eisenberg,  New  Broadcast  Program  Transmission  Standards,  JFYac,  1.&JR*  1M7. 
Chinn,  EL  A.,  and  P.  Eisenberg,  Influence  of  Reproducing  System  on  ToaaJ-jaaage  PiBfereaeas,  Prec. 

I.R.E.,  May  1948,  p.  572. 
Gannett,  D.  EL,  and  I.  Kerney,  Discernibility  of  (Granges  in  Program  Band  Width,  B*&  S&s.  fadt,  J., 

January  1944,  p.  1. 
Seacord,  D.  F.,  Room  Noise  at  Telephone  Locations,  Eiec,  B*x~,  Jtuae  1939,  p.  225;  June  194O,  p.  232. 


POLICE  RADIO 

By  H.  F.  Micfcd 

Police  radio  systems  may  be  divided  into  two  major  elassificaiioiis  from  an  operational 
standpoint.  The  simplest  type  of  system,  designated  "oae  way/'  permits  communica- 
tion in  one  direction  only,  from  the  headquarters  station  to  mobile  units.  This  type  of 
operation  requires  a  land  station  transmitter  for  the  headquarters  location  and  a  receiver 
for  each  mobile  unit.  The  "two-way"  system  provides  communication  from  the  iiead- 
quarters  station  to  mobile  units  and  from  mobile  imits  to  headquarters.  The  equrpsaeat 
required  for  a  two-way  system  consists  essentially  of  a  land  station  taasmittec,  one  or 
more  land  station  receivers,  and  a  mobile  receiver  and  transmitter  for  each  two-way  ve- 
hicle. In  certain  installations  where  all  equipment  is  oa  the  same  freqvteney,  or  wliere 
mobile  transmitters  are  equipped  for  1rfro-freqiien.cy  operation,  a  " three-way"  system  is 
evolved  permitting  car-to-car  communicatioii  in  addition  to  the  two-way  previously  de- 
scribed. Essentially  all  systems  which  have  been  placed  in  service  siace  1942  are  of  tfae 
two-way  or  three-way  type. 

Police  radio  systems  may  also  be  divided  into  two  major  group©  on  tfce  basis  of  the 
kind  of  equipment  used.  Prior  to  1940,  practically  all  police  radio  installations  employed 
a-m  apparatus.  Since  that  date,  the  vast  majority  of  systems  have  made  use  of  f-ua 
equipment.  The  only  activity  in  the  installation  of  a-m  apparatus  is  confined  to  the  re- 
placement or  expansion  of  existing  systems.  New  systems,  almost  without  exception,  are 
of  the  f-m  variety. 

The  scope  and  complexity  of  police  radio  systems  vary  with  the  requirements  of  eaeii 
particular  installation.  A  small  municipality  may  operate  a  single  headquarters  station 
and  a  small  number  of  mobile  units.  If  conditions  at  the  system  control  point  are  ao* 
desirable  for  the  local  installation  and  control  of  the  land  station,  a  location  providing 
advantages  of  increased  elevation  and  improved  noise-level  conditions  may  be  selected  for 
this  equipment.  This  requires  remote-control  apparatus  at  the  control  point  and  the 
interconnection  of  the  control  and  station  locations  by  means  of  wire  line. 


16-36  SOUND-REPRODUCTION  SYSTEMS 

In  larger  cities  where  a  great  number  of  mobile  units  and  a  considerable  coverage  area 
are  involved,  fixed  station  equipments  may  be  located  at  several  points  with  individual 
control  from  one  central  dispatching  station  or  from  separate  precinct  control  points. 

In  many  cases,  individual  receivers  of  the  type  used  in  land  stations  are  located  at 
several  advantageous  points  throughout  a  city  with  their  outputs  feeding  back  to  the 
control  station  or  stations  over  wire  line.  This  greatly  increases  the  talk-back  range  of 
the  mobile  unit  to  the  fixed  station. 

State  police  systems  normally  involve  a  multiplicity  of  land  station  transmitters  and 
receivers  located  strategically  to  cover  desired  troop  or  patrol  areas. 

A  number  of  state  and  large  city  systems  also  incorporate  CW  telegraph  stations  for 
zone  and  interzone  point-to-point  communication. 

In  some  instances,  radio  relay  equipment  is  used  for  the  control  of  remotely  located 
stations,  particularly  where  topographical  conditions  necessitate  such  remote  installation 
of  land  station  equipment  and  render  impractical  the  use  of  wire  line  interconnection  to 
the  control  station. 

19.  FREQUENCIES 

The  first  police  radio  systems  operated  on  frequencies  just  above  the  standard  broad- 
cast band  with  channels  in  two  portions  of  the  spectrum  (1610  kc-1730  kc  and  2326  kc- 
2490  kc)  assigned  for  this  purpose.  The  general  plan  was  to  place  state  systems  in  the 
lower  band  and  city  stations  on  the  higher  channels.  A  number  of  these  systems  are  still 
in  operation,  and  essentially  all  apparatus  employing  these  frequencies  is  for  land-station- 
to-mobile  communication.  These,  doubtless,  will  be  gradually  replaced  by  equipment 
operating  on  higher  frequencies. 

The  next  portion  of  the  spectrum  assigned  for  police  operation  was  the  30.1-40.1  mega- 
cycle band  (now  30—50  megacycles) .  It  was  found  that  these  frequencies  possessed  many 
operational  advantages  for  police  work:  reduction  of  interference  between  stations,  re- 
duced atmospheric  and  man-made  interference,  lower  transmitter  power  output  require- 
ments, and,  probably  most  important  of  all,  the  ability  to  produce  practical  equipment  for 
mobile  talk-back  operation.  It  was  in  this  band  that  f-m  apparatus  for  mobile  com- 
munications made  its  appearance  with  its  many  attendant  advantages. 

During  World  War  II  the  use  of  still  higher  frequencies  by  the  Armed  Forces  disclosed 
many  characteristics  which  pointed  toward  the  desirability  of  their  use  for  police  communi- 
cations systems.  Some  background  of  experience  gained  in  the  use  of  116-  to  118-mega- 
cycle  equipment  for  relay  purposes  in  police  systems,  starting  about  1940,  also  gave  added 
weight  to  this  belief.  Experimental  work  encompassing  frequencies  in  the  160-megacycle 
area  revealed  very  favorable  performance  characteristics  for  police  use.  Accordingly, 
the  Federal  Communications  Commission  has  set  aside  channels  in  the  152-  to  162-mega- 
cycle  band  for  the  police  services.  From  actual  data  covering  a  representative  number  of 
systems  in  normal  operation,  it  appears  that  sky-wave  or  "skip"  interference,  often  caus- 
ing considerable  trouble  in  the  30-  to  50-megacycle  band,  is  greatly  reduced  in  the  152- 
to  162-megacycle  frequencies.  Smaller  antennas  are  a  natural  and  desirable  result  of  the 
use  of  152-  to  162-megacycle  channels  with  the  further  advantage  that  high  gain  and 
directional  antennas  for  land  stations  are  entirely  feasible  at  these  frequencies.  Atmos- 
pheric and  man-made  interference  is  still  further  reduced  as  compared  to  the  30-  to  50- 
megacycle  band.  Coverage  within  the  normal  service  area  of  a  152-  to  162-megacycle  sys- 
tem is  more  complete,  with  fewer  dead  spots,  than  with  lower  frequencies.  It  appears,  also, 
that  still  lower  transmitter  powers  will  give  satisfactory  results. 

The  Federal  Communications  Commission  has  also  made  certain  shared  assignments  for 
police  service  in  the  72-  to  76-megacycle  band.  Tests  conducted  on  the  30-  to  50-mega- 
cycle, 72-  to  76-megacycle,  and  the  152-  to  162-megacycle  bands  indicate  that  the  extreme 
coverage  range  decreases  somewhat  as  the  frequency  rises  but  that  improved  blanket 
coverage  within  the  useful  range  is  achieved  as  the  frequency  is  increased.  The  30-  to  50- 
megacycle  channels  are  characterized  by  a  greater  degree  of  "bending"  of  the  transmitted 
signal  and,  for  that  reason,  seem  better  suited  for  applications  where  greater  coverage 
distance  is  required,  particularly  in  hilly  or  mountainous  terrain. 

It  appears,  therefore,  that  30-  to  50-megacycle  channels  are  best  suited  for  state  and 
county  police  systems  and  152-  to  162-megacycle  frequencies  for  municipal  use.  The 
Federal  Communications  Commission  has  also  provided  bands  for  police  service  in  various 
portions  of  the  spectrum  from  450  megacycles  to  30,000  megacycles.  Complete  informa- 
tion regarding  all  allocations  of  frequencies  for  police  use  may  be  obtained  from  the  May  6, 
1949,  issue  of  the  Federal  Register  or  from  the  new  FCC  Rules  and  Regulations,  when 
published. 


EQUIPMENT 


16-37 


20.  POWER  AND  RANGE 

There  is  no  fixed  formula  for  the  absolute  determination  of  coverage  which  may  be 
expected  in  a  police  radio  system.  Local  conditions  of  terrain,  antenna  elevation,  and 
noise  level  are  some  of  the  variables  that  influence  such  coverage.  Transmitter  power 
output  is  also  a  factor,  particularly  in  the  1610-  to  1730-kc  and  2326-  to  2490-kc  bands. 
However,^  in  the  higher-frequency  channels,  antenna  elevation  and  noise-level  conditions 
are  more  influential  than  transmitter  power.  Table  1  indicates  the  normal  limit  of  trans- 
mitter power  in  the  various  frequency  bands  (plate  power  input  to  final  stage). 

Since  communications  range  is  a  function  of  so  many  variable  factors,  actual  experience 
in  the  planning  and  installation  of  police  radio  systems  is  the  most  reliable  means  of  pre- 
dicting results.  A  single  land  station  installation  will  afford  satisfactory  two-way  communi- 
cation for  an  average  city  county,  provided  that  the  antenna  site  is  carefully  selected 
from  the  standpoint  of  elevation  and  noise  level.  Two-way  range  up  to  50  miles  with 
30-  to  50-megacycle  equipment  is  not  uncommon  with  modern  apparatus  properly  in- 
stalled. A  slight  decrease  in  range  may  be  expected  with  72-  to  78-iDegacydte  and  152-  to 
162-megacycle  equipment. 

Table  1 


Frequency, 
megacycles 

Land  Station 
Power, 
watts 

1.61-3.0 
25-100 
100-220 
Above  220 

2060 
50© 

&m 

To  be  specified 
in  authorisation 

21.  EQUIPMENT 

Land  station  transmitting  equipment  is  of  conventional  design  using  standard  circuits 
and  tubes.  F-m  equipment  is  normally  of  the  phase-shift  type.  The  same  applies  to 
mobile  transmitters. 

Land  station  and  mobile  receivers  are  of  the  superheterodyne  variety  and  are  filed 
tuned  to  the  assigned  operating  frequency.  Squelch  circuits  are  provided  to  quiet  the 
receivers  when  the  associated  carrier  is  not  on  the  air. 

Crystal  control  has  become  standard  for  all  transmitting  and  receiving  equipment  de- 
signed for  police  service. 

Power  for  land  station  equipment  is  normally  obtained  from  the  regular  public  utility 
service.  Frequently  gas-engine  generating  equipment  is  provided  for  emergency  opera- 
tion in  the  event  of  failure  of  the  regular  power  source. 

Mobile  equipment  uses  the  car  battery  as  the  primary  power  source  with  either  vibrator 
or  dynamotor  units  for  high-voltage  d-c  supply. 

Since  vertical  polarization  has  become  standard  in  police  service  (30  megacycles  and 
up),  mobile  antennas  are  of  the  vertical  whip  type  for  either  side  or  roof  top  mounting, 
Land  station  antennas  vary  somewhat  in  design  but  are  usually  of  the  J,  coaxial,  or  ground 
plane  type. 


SECTION  17 
TELEPHONY 


JOHN  D.  TAYLOR 


J^Y      CENTRAL-OFFICE  EQUIPMENT  PAGK 

L  Manual  Systems  and  Operation 03 

2.  Mechanical  Systems  and  Operation 08 

3.  Toll  Systems  and  Operation 36 

4.  Tandem  Systems  and  Operation 49 

5.  Auxiliary  Service  Equipment 51 

6.  Common.  Systems. 51 

7.  Power  Systems. , 53 

RADIO  TELEPHONE  SYSTEMS 

8.  Applications. 55 

9.  Transmission  and  Operational  Methods.  61 

10.  Principles  of  Two-way  Operation 62 

11.  System  Design 64 

12.  Installations 66 


TELEPHONE  LINES— TRANSMISSION  CON- 
ART  SIDERATIONS  PAOE 

13.  Types  of  Plant , 69 

14.  Service  Requirements— Toll 69 

15.  Service  Requirements — ExeJmnge.  .....     78 

16.  Plant  Design — Toll , .     82 

17.  Flaat  Design — Exefa&age 92 

PROGRAM  SERVICE 

18.  Program  Service. 


101 


SOBSCKIBEB.  STATIONS 

19.  Substation  Eqaipraeafc, 106 

20.  Subscriber  Station  Protection 113 

21.  Private  Branch  Exehaage  Equipment . . .   114 


17-01 


LOCAL  OFFICE 


TELEPHONY 

By  John  D.  Taylor 

TELEPHONY  is  the  art  of  electrically  transmitting  speech  between  two  01  more  points. 
Telephone  facilities  are  also  used  for  many  other  purposes,  such  as  the  transmission  of 

broadcasting  and  public-address  programs. 
Transmission  of  speech  and  other  forms  of 
intelligence  is  accomplished  over  wire  circuits 
or  through  the  air  (radio)  or  by  a  combina- 
tion of  both  mediums. 

The  telephone  circuit  fundamentally  con- 
si3*8  °f  a  Device  (transmitter)  for  transform- 

«  •,  j        •      ,  i        ,     •       i 

**  *>  fch  SOimds  ^°  electrical  currents, 
which  traverse  a  connecting  medium  (line  or 
channel)  and  react  in  another  device  (receiver) 


SUBSET 


LOOP 


FIG.   1.     Connection  between  Two  Subscribers 
in  the  Same  Office  (Courtesy  Bell  System) 


LOOP 


LOOP 


in  such  manner  as  to  convert  the  electrical  currents  into  the  original  speech  sounds. 

Switching  arrangements  of  various  types  and  capacities,  either  manual  or  mechanical,* 
are  necessary  to  connect  local  or  toll  telephone  circuits  together,  and  a  number  of  auxiliary 
circuits,  in  addition  to  the 

talking  circuit,  may  be  em-  LOCAL  OFFICE  LOCAL  OFFICE 

ployed  for  a  given  connec-     SUBSET  I  I  SUBSET 

tion,  depending  upon  the 
types  of  systems  involved 
and  the  length  of  the  con- 
nection. The  interconnec- 
tion of  two  subscriber  lines 
(loops)  in  the  same  office  is 
usually  quite  simple,  but  for 
lines  in  widely  separated 
offices  the  complete  interconnecting  circuit  and  associated  apparatus  may  be  very  complex. 

Representative  types  of  telephone  connections  between  two  subscribers  are  shown 
schematically  in  Figs.  1,  2,  3,  and  4. 


MANUCONNECTiONANICAL 

2     Connection  between  Two  Subscribers  in  Different  Offices 
^  the  Same  City  (Courtesy  Bell  System) 


OFFICES 


if 

LOCAL 


TOLL 


LOCAL 


UBSET 

*  TOLL 
CONNECT- 
ING 
TRUNK 

-«_o- 

TOLL 
CIRCUIT 

-y- 

TOLL 
SWITCH 
TRUNK 

*»• 

-y 

/ 

LOOP 

\ 

\ 

\ 

/ 

SUBSET 


LOOP 


^-_«.  MANUAL  OR  MECHANICA1 , 

CONNECTION- 

FIG.  3.     Connection  between  Two  Subscribers  in  Different  Cities  (Courtesy  Bell  System) 


5UBSE* 

TC 

r 

)LL  OFFIC 

E 

TRANS- 
MITTING 

r~ 
i 

1 

! 

i_ 

RE- 
CEIVING 

THROUGH     INTERMEDIATE 
OFFICES 


RADIO     TERMINALS 


RE- 
CEIVING 

T< 

JLL   OhHC 

6 

4-.~_- 

| 

/ 

TRANS- 
MITTING 

SUBSET 


THROUGH    INTERMEDIATE 
OFFICES 


PIG.  4.     Connection  between  Two  Subscribers  in  Different  Countries  via  Radio  Channel  (Courtesy 

Bell  System) 

*  The  word  "mechanical"  is  used  in  this  section  in  a  broad  sense  to  include  all  forms  of  non-manual 
switching. 

17-02 


MANUAL  SYSTEMS  AND  OPERATION  17-03 

In  telephone  practice  the  various  facilities  naturally  fall  under  four  main  headings: 
(1)  central-office  equipment;  (2)  land  and  buildings;  (3)  telephone  lines;  (4)  substation 
eauroment. 


CENTRAL-OFFICE  EQUIPMENT 


^  Central-office  equipment,  in  general,  embraces  the  various  switching  arrangements, 
including  auxiliary  units  of  equipment,  which  are  necessary  for  the  interconnection,  dis- 
connection, control,  and  super-vision  of  telephone  facilities.  Usually,  the  larger  the 
telephone  system,  the  more  intricate  are  the  equipment  requirements. 

The  evolution  of  telephone  service  has  been  from  magneto  to  manual  common-battery 
to  mechanical  Common-battery  operation.  All  these  types  of  operation  are  now  in  use, 
but  the  trend  in  present-day  engineering  is  to  mechanize  telephone  service  in  order  to  ob- 
tain greater  speed,  ease,  and  efficiency  of  operation  and  to  avoid  higher  operating  costs. 

Central-office  equipment  includes  all  types  of  switchboards,  both  manual  and  mechan- 
ical,^ switch  frames  and  panels,  terminating  frames  and  racks,  toll  terminal  equipment, 
testing  units,  power  plants,  and  many  auxiliary  pieces  of  apparatus.  The  equipment  is 
housed  in  suitable  buildings,  and  each  entire  assembly  is  known  as  a  central  office. 

Auxiliary  circuits  and  apparatus,  such  as  alarms  and  indicators,  both  visual  and  aural, 
designed  to  call  attention  to  certain  operating  conditions,  monitoring  and  supervisory 
circuits,  timing  and  recording  devices,  emergency  power  and  ringing  circuits,  test  circuits, 
and  many  other  devices,  necessary  for  the  proper  operation  of  central  offices,  are  common 
to  all  systems  to  a  greater  or  less  degree,  depending  on  the  type  and  size  of  system. 

1.  MANUAL  SYSTEMS  AND  OPERATION 

Manual  systems  include  both  magneto  and  common-battery  systems,  in  which  tele- 
phone operators  manually  establish  and  supervise  connections  at  switchboards,  using 
flexible  cords  or  keying  units. 

Magneto  operation,  first  employed  in  the  United  States,  requires  operation  of  a  magneto 
or  hand  generator  (associated  with  magneto  telephone  sets)  by  the  subscriber  to  signal  the 
operator  for  connections  and  disconnections,  and  the  provision  of  local  battery  (dry  cells* 
at  each  telephone.  Present  practice  also  provides  for  hookswiteh  signaling  (with  limita- 
tions) by  the  subscriber  on  magneto  lines,  if  desired,  similar  to  common-battery  operation. 

Common-battery  operation  provides  for  the  subscriber  to  signal  the  operator  by  re- 
moving his  handset  *  from  or  replacing  it  on  the  telephone  set  hook,*  direct  current  for 
both  signaling  and  talking  being  supplied  to  the  telephone  set  from  the  central-office  bat- 
tery over  the  subscriber  line. 

Manual  switchboards  are  of  several  types  and  are  made  by  a  number  of  different  manu- 
facturers for  inter  connecting  toll,  trunk,  and  subscriber  line  circuits. 

Magneto  switchboards  employ  simple  cord  and  line  circuits  but,  in  general,  provide 
the  least  desirable  telephone  service  from  the  standpoint  of  speed  and  ease  of  operation. 
These  boards  are  now  built  with  capacities  of  up  to  200  subscriber  Hues  and  are  used 
principally  in  small  offices,  where  the  majority  of  the  terminating  lines  extend  into  rural 
areas  and  have  relatively  high  energy  losses.  Even  here  the  present  tendency  is  toward 
mechanical  equipment  in  new  installations  and  replacements. 

Typical  full  magneto  switchboard  circuits  of  the  latest  type,  for  a  board  having  a  capacity 
of  150  lines  and  15  cord  circuits,  are  shown  in  Fig.  1. 

FULL  MAGNETO  OPERATION  of  such  a  switchboard  is  as  follows: 

In  placing  a  call,  the  subscriber  turns  the  hand  generator  crank  at  his  telephone,  send- 
ing 20-cycle  current  over  his  line  and  operating  the  switchboard  drop  (or  line  lamp  cir- 
cuit, if  furnished).  The  subscriber  then  removes  his  handset  from  the  hook  and  listens 
for  the  operator  to  answer.  The  operator  inserts  the  answering  plug  of  an  idle  cord  cir- 
cuit in  the  line  jack  associated  with  the  operated  drop,  opens  her  listening  key,  and  requests 
the  called  number.  Assuming  that  this  number  is  also  a  local  subscriber's  line  in  this 
office,  the  operator  inserts  the  calling  plug  (associated  with  the  cord  circuit  being  used)  in 
the  called-number  line  jack  and  operates  her  ringing  key,  using  code  ringing  as  may  be 
required,  to  signal  the  called  station.  If  ringing  power  is  supplied  to  the  switchboard  by 
a  hand  generator,  the  generator  crank  must  also  be  turned  by  the  operator  while  operating 
the  ringing  key.  The  called  station  bell  is  actuated  by  the  20-cycle  current  sent  out  over 

*  The  word  "handset"  is  intended  to  include  the  older-type  telephone  receiver  and  the  word  "hook" 
to  include  the  newer-type  telephone-set  cradle. 


17-04 


TELEPHONY 


the  line,  and  the  operator  awaits  the  called  subscriber's  answer  with  her  listening  key  open; 
when  she  hears  the  subscriber  answer  she  closes  her  key  and  disconnects  her  telephone  set 
from  the  connection. 

On  completion  of  the  call  both  subscribers  place  their  handsets  on  the  hook,  the  calling 
subscriber  turns  his  generator  crank,  operating  the  answering  cord  ring-off  drop  and  sig- 


(a) 


|r5oo*n 

R 

r* 

DV  *     o 

TO  UNE 

**-T~ 

T 

D  FRAME                                                                              I 

REPEATING  COIL 
CUT-OUT  KEY 


(b) 


v  BATTERY      KEY,        TO   KEY 
TO  NIGHT  ALARM    CODE  ALARM 


OPERATOR'S 
TELEPHONE 


(c) 


INDUCTION 

COI 

L 

c 

^ 

3       C 

1 

[^ 

11 

i  j 

CONTACTS    ON           1 
LISTENING  KEY 

-<TRnr 

4-t 

•v 

A 

jf  

* 

MONITOR 
REPEATING 
COIL 
i 

VARiSJOR  j 

^ 

°     4 

GROUPING  KEY 
FOR  POSITION 
SWITCHING 

|  MONITOR  KEY  ^^IJ   „ 

U&b 

L-h/   flS 

<")~~A                                  ! 

1-ffffflP            i  }  1  

R!    IT 

TO  LISTENING  KEY 

FIG.  1.    Full  Magneto  Switchboard  Circuits— Magneto  Signaling  by  Subscriber  (Courtesy  Stromberg- 

Carlson  Co.) 

(a)  Line  circuit  with  drop  arranged  for  code  alarm. 

(6)  Cord  circuit^with  repeating  coil,  double  clear-out  drops,  and  repeating  coil  cut-out  key. 
(c)   Operator  s  circuit,  including  monitor,  varistor,  and  grouping  key  circuits. 

naling  the  operator  that  the  conversation  is  completed.     She  then  removes  the  plugs  from 
the  line  jacks  and  restores  the  cord  circuit  to  its  idle  position. 

Central  energy  (common-battery)  type  telephones  on  magneto  lines  may  be  made  to 
operate  successfully,  using  a  line  circuit,  as  shown  in  Fig.  2.  This  circuit  provides  service 
to  a  subscriber  in  a  magneto  office  similar  to  common-battery  operation,  but  such  service 
is  limited  generally  to  short  town  lines,  not  exceeding  about  225  ohms  conductor  loop  re- 
sistance. The  subscriber  removes  his  handset  from  the  hook,  sending  a  surge  of  direct 
current  through  the  repeating  coil  windings  (line  side},  the  line,  and  the  telephone  set. 
This  surge  induces  a  surge  of  current  in  the  drop  windings  of  the  repeating  coil,  which  are 
in  series  with  the  line  drop.  The  line  drop  is  operated  and  the  caU  is  handled  from  that 
point  in  the  same  manner  as  other  full  magneto  calls. 


MANUAL  SYSTEMS  AND  OPERATION 


17-05 


Magneto  toll  lines  are  terminated  at  magneto  switchboards  in  the  same  type  of  line 
circuit,  and  the  operating  is  similar  to  that  for  local  circuits.  In  the  local  switchboard, 
two  pairs  of  cord  circuits  are  arranged  so  that  the  repeating  coil  can  be  removed  by  operat- 
ing a  key  on  toll  connections,  in  order  to  reduce  transmission  loss,  assuming  that  circuit 


REPEATING   COIL 
5 


CENTRAL  OFFICE 
PROTECTION 


FIG.  2.     Magneto  Switchboard  Subscriber  line  Circuit  Arranged  for  Common-battery  Signaling  aaad 
Talking  by  Subscriber  i,Courtesy  Stromberg-Carlson  Co.) 

noise  is  satisfactory  without  the  coil  for  a  particular  connection.  Figure  3  shows  typieal 
loop  and  simplex  dial  trunk  and  cord  circuits  for  use  &t  magneto  switchboards,  which 
have  trunks  to  a  mechanical  office. 


WIPE-OUT  KEY 


DIAL  CORD 

emeu  IT 

IMF 

\{ 

C*    °500*»JNOH-            ^ 
")  INDUCTIVE 

H 

1 

RETARDATION 
COIL 


REPEATING 


CHAL  JACK 

LOOP  DIAL  TRUIviK  CIRCUIT 


JACK      -=- 

S1MPLEX  DIAL  TRUNK  CIRCUIT 


NOTE  A!    LOOP  DIAL  MAY  fiE  CHANGED  TO  SIMPLEX  DIAL  TRUNK   CIRCUITS   &f  ADDING  REPEATING  COIL, 
DISCONNECTING  RETARDATION  COJL,  AND  MAKING  PROPER   CONNECTIONS. 

NOTE  8*.    SIMPLEX    DIAL   MAY  BE  CHANGED  TO  LOOP  DIAL  TRUNK  CIRCUITS  Efcf  ADDING  RETARDATWN 
COM.,  DISCONNECTING  REPEATING  COIL,  AND  MAKING  PROPER  COMN£CDO*4S. 

TIG.  3.     Magneto  Switchboard— Dial  Trunk  and  Cord  Circuits  (Courtesy  Siromberg-CarJsoB  Co.) 

COMMON-BATTERY  SWITCHBOARDS  are  made  in  a  variety  of  types  and  capaci- 
ties, both  single  and  multisection  (multiple),  to  meet  service  requirements  a&d  are  widely 
used  throughout  the  United  States  and  foreign  countries.  The  subscriber  signals  the 
switchboard  operator  by  operating  the  hookswitch  (or  cradleswitch)  of  his  telephone. 
However,  in  order  to  provide  this  more  convenient  and  faster  service,  the  subscriber  switch- 
board line,  cord,  and  auxiliary  circuits  are  more  complex  than  for  magneto  equipment. 


17-06 


TELEPHONY 


Single-section  common-battery  non-multiple  switchboards  are  used  principally  in  the 
smaller  towns,  where  magneto  service  is  not  adequate.  In  this  type  of  board  each  sub- 
scriber's line  appears  in  the  switchboard  jack  field  only  once,  since  one  operator  can  reach 


any  jack  in  the  board.  Figure  4  shows  schematic  circuits  of  a  typical  board  of  this  type. 
The  capacity  of  such  a  board  is  up  to  200  subscriber  lines,  30  toU  or  rural  lines,  and  16 
universal  cord  circuits.  The  capacity  may  be  doubled  by  operating  two  such  boards  ad- 
jacent to  each  other. 


MANUAL  SYSTEMS  AND  OPERATION 


17-07 


Multiple  common-battery  switchboards  are  designed  for  single-office  and 
cities,  where  single-section  boards  are  inadequate  to  meet  service  requirements.  The  ca- 
pacities of  this  type  of  board  range  from  about  600  to  about  10,000  lines,  thus  limiting  the 
capacity  of  a  single  office  to  about  10,000  lines.  For  the  large  cities,  requiring  more  than 
10,000  lines,  more  than  one  office,  each  with  its  own  switchboard,  is  necessary. 

Single-office  multiple  common-battery  switchboards  are  assembled  by  sections  in  one 
or  more  line-ups,  each  section  being  identically  equipped  with  jack  Selds  and  cord  circuits. 
The  number  of  sections  in  an  office  varies  from  two  to  twenty  or  more,  depending  upon 


TO  OTHER 
MULTIPLE       ^ 
JACKS  AND  ] 
LAMPS 


TO  SUPERVISORY  LAMP  CIRCUIT 
SUBSCRIBER   Oft  RURAL  CORD  CIRCUIT 


—  ~ ) 


INDUCTION   COR. 


REPEATING 

cot. 


OPERATOR'S  TELEPHONE 


FIG.  5.    Multisectaon  Common-battery  Switchboard  Circuits  —  4000-Hae  Commota-battery  Talking  »od 
Signaling  (Courtesy  Bell  System) 

the  number  of  subscriber  lines  served  and  the  traffic  load.  Each  section  of  switchboard 
provides  for  from  one  to  two  and  two-thirds  operators  and  from  three  to  eight  panels,  in 
which  the  multiple  jack  and  lamp  strips  are  mounted.  Each  subscriber  line  has  oae 
multiple  jack  with  associated  line  lamp  in  each  section,  although  in  some  of  the  older 
boards  only  one  answering  jack  was  provided  per  line.  Thus,  jack  100  and  its  associated 
lamp  in  the  first  section  in  the  line-up  are  cabled  to  jack  100  and  its  lamp  in  the  second  sec- 
tion, and  similarly  throughout  the  board. 

Since  each  subscriber  line  terminates  in  the  jack  and  lamp  circuit  corresponding  to  his 
number,  when  the  subscriber  signals  the  operator  to  place  a  call  all  the  line  lamps  associated 
with  his  line  throughout  the  board  light  and  may  be  answered  by  any  available  operator, 
but  by  only  one  at  a  time.  When  one  operator  answers  a  call,  all  jack  sleeves  associated 
with  that  particular  line  have  potential  placed  on  them*  which  causes  a  click  in  the  ear  of 


17-08  TELEPHONY 

any  operator  who  touches  the  tip  of  her  plug  to  the  sleeve  of  any  jack  associated  with  that 
line.  This  click  warns  the  operator  that  the  line  is  busy.  The  number  of  line  lamps  which 
are  permitted  to  light  on  any  one  line  is  usually  limited  to  five  but  may  be  less,  depending 
on  traffic  loads  and  calling  rates.  Figure  5  shows  the  principal  schematic  circuits  for  a 
typical  multiple  common-battery  switchboard  with  a  capacity  of  4000  subscriber  lines, 
360  toll  lines  or  720  outgoing  trunks,  and  17  cord  circuits  per  position, 

The  provision  of  a  trunking  board  and  special  arrangements  of  subscriber  multiple  in 
the  various  line-ups  makes  it  possible  to  increase  the  capacity  of  the  board  to  accommodate 
up  to  5600,  7200,  or  10,400  subscriber  lines  and  to  provide  for  a  substantial  complement  of 
toll  lines  and  trunks.  However,  when  new  central-office  installations  or  sizable  additions 
to  existing  manual  boards  are  being  considered,  present  practices  require  a  careful  study 
to  determine  the  practicability  and  economies  of  employing  mechanical  operation,  be- 
cause of  its  many  advantages,  including  integration  with  the  general  trend  toward  uni- 
versal mechanized  telephone  service. 

This  board  is  capable  of  operating  as  a  combined  local  and  toll,  local  and  trunk,  or  local, 
toll,  and  trunk  board. 

Multioffice  multiple  common-battery  switchboards  of  modern  design  are  similar  to  the 
single-office  multiple  board  described  above.  Some  of  the  older-type  subscriber  switch- 
boards did  not  provide  for  multipling  the  line  lamp  as  well  as  the  multiple  jack,  so  that 
the  subscriber's  lamp  signal  had  only  one  appearance  in  the  entire  switchboard  and  answer- 
ing time  was  considerably  slower  than  with  the  multiple-line  lamp  arrangement. 

INTEROFFICE  TRUNKS  are  necessary  in  multioffice  exchange  areas  to  provide  for 
extending  a  call  from  one  office  to  another.  In  manual  operation  the  calling-subscriber 
signal  appears  at  the  calling-subscriber  switchboard  (designated  in  trunking  as  the  A 
board) ,  the  operator  ascertains  the  called  number  and,  either  by  the  call  circuit  or  straight- 
forward trunking  method,  passes  the  called  number  to  a  terminating  trunk  board  (desig- 
nated in  trunking  as  the  B  board)  at  the  called  office.  An  intermediate  office  (tandem) 
may  be  involved  in  establishing  the  trunk  connection.  The  A  operator  connects  the 
calling  line  to  the  selected  outgoing  trunk  at  the  A  board,  the  B  operator  at  the  called 
office  connects  the  B  end  of  the  interconnecting  AB  trunk  to  the  called  B  board  multiple 
jack,  and  the  ringing  of  the  called  subscriber  automatically  starts.  The  cycle  of  ringing 
usually  consists  of  a  2-sec  ringing  interval  followed  by  a  4-sec  silent  interval  with  d-c 
potential  only  impressed  on  the  line.  This  cycle  is  repeated  until  the  subscriber  answers 
or  the  connection  is  taken  down.  When  the  subscriber  answers  in  either  the  ringing  or 
silent  interval,  relays  in  the  B  trunk  circuit  operate,  disconnecting  the  ringing  power 
and  connecting  the  trunk  circuit  talking  path  through  to  the  called  subscriber.  Upon 
completion  of  the  conversation,  a  lamp  disconnect  signal  appears  before  both  A  and  B 
operators,  in  response  to  both  the  calling  and  called  subscribers  hanging  up  their  hand- 
sets, and  the  connection  is  taken  down. 

Call  circuit  trunking  is  a,  procedure  by  which  the  A  or  toll  operator  passes  a  call  to  a 
B  (or  tandem)  operator  over  a  call  (order  wire)  circuit,  which  is  entirely  separate  from  the 
trunk  circuit  being  used  for  the  call.  When  the  A  operator  presses  her  call  circuit  key 
associated  with  the  call  circuit  to  the  desired  B  (or  tandem)  office,  she  is  connected  directly 
to  the  distant  operator's  telephone  set.  After  she  passes  the  call  to  the  distant  operator, 
the  distant  operator  assigns  an  idle  trunk  in  the  group  between  the  two  offices  and  the  A 
and  distant  operators  connect  the  trunk  to  the  calling  and  called  lines  at  their  respective 
boards. 

Straightforward  trunking  is  now  the  generally  accepted  method  in  manual  operation 
rather  than  the  call  circuit  method,  from  which  it  differs  in  that  the  A  operator  selects  the 
idle  trunk  to  the  called  office.  She  then  connects  the  calling  subscriber  to  the  trunk  with 
an  A  cord  circuit,  causing  a  lamp  to  light  at  the  distant  operator's  position.  The  distant 
operator  connects  her  telephone  set  to  the  trunk  by  pressing  a  key,  or  the  set  is  automati- 
cally connected  and  the  A  operator  is  so  informed  by  hearing  a  two-tone  signal  on  the 
trunk.  The  call  is  passed  and  the  connection  is  established,  and  during  the  conversation 
the  supervisory  lamps  at  both  A  and  B  boards  remain  dark.  When  the  subscribers  hang 
up  their  handsets  these  lamps  light  and  the  operators  disconnect. 

2.  MECHANICAL  SYSTEMS  AND  OPERATION 

Of  the  several  types  of  mechanical  switching  systems  now  in  operation,  probably  the 
Strowger  (step-by-step)  system,  manufactured  by  the  Automatic  Electric  Co.  (and  others 
under  Automatic  Co.  patents)  is  most  widely  known.  It  was  the  first  type  employed  com- 
mercially (in  the  year  1892)  and  is  still  used  extensively  today.  Other  well-known  mechan- 
ical systems  have  been  developed  to  meet  the  needs  of  the  rapidly  growing  telephone  in- 


MECHANICAL  SYSTEMS  AND  OPERATION 


17-09 


dustry,  particularly  the  Relaydiai  (Stromberg-Carleoii),  Relaymatic  (Kellogg  Switchboard 
and  Supply  Co.),  All-Relay  (North  Electric  Manufacturing  Co.),  Panel  Dial,  and  Crossbar 


Systems  (Western  Electric  Co.).  AH  these  systems  have  only  one  purpose,  namely,  to 
switch  traffic  quickly,  accurately,  economically,  and  in  a  manner  satis! actory  to  the  public, 
whether  it  be  in  a  relatively  small  office  or  in  the  largest  multioffice  exchange  area- 


17-10 


TELEPHONY 


THE  STRO WGER  SYSTEM  employs  the  well-known  step-by-step  method  of  operation, 

so  called  because  calls  are 
advanced  from  the  calling 
to  the  called  subscriber  step 
by  step,  as  each  digit  of  the 
called  number  is  dialed  by 
the  calling  subscriber. 

In  step-by-step  operation, 
the  principal  switching  units 
involved  in  a  connection  be- 
tween two  local  subscribers 
are  a  line  switch  or  line 
finder,  one  or  several  ranks 
of  selectors,  and  a  connec- 
tor. In  addition,  if  the 
connection  includes  a  trunk 
between  two  offices  in  the 
same  exchange  area,  the 
outgoing  end  of  the  trunk 
will  terminate  in  an  im- 
pulse repeater. 

The  equipment  which 
appears  between  the  sub- 


FIG.  7. 


Self-aligning  Plunger  Line   Switch 
Electric  Co.) 


(Courtesy   Automatic 


scriber's  line  terminals  and  the  first  rank  of  selectors  is  classed  as  non-numerical,  since  it 
automatically  functions  as  soon  as  the  subscriber's  handset  is  removed  from  its  support 
and  before  any  digits  are  dialed. 
Non-numerical  switches  are  of  two 
major  classes — line  switches  and  line 
finders.  The  line  switch  is  individ- 
ual to  a  telephone  line  and  serves  to 
extend  the  calling  line  to  an  idle  se- 
lector or  connector  (forward  selec- 
tion) ,  while  a  line  finder  is  connected 
permanently  to  a  selector  or  connec- 
tor and  serves  to  find  the  calling  line 
(backward  selection) .  Line  switches 
are  now  seldom  employed  for  public 
exchanges  but  are  generally  standard 
for  private  automatic  exchanges  of 
100  lines  or  less. 

The  line  switch  may  be  of  two 
types,  plunger  (10-trunk  capacity) 
and  rotary  (10-  or  25-trunk  capacity). 

The  plunger  line  switch  is  a  simple 
mechanism  which  automatically  con- 
nects  its  calling  Line  to  any  one  of  a 
number  of  trunks  leading  to  numeri- 
cal switches  (the  selector  or  connec- 
tor, which  are  operated  by  dial 
pulses) .  Figure  6  shows  a  schematic 
diagram  of  the  self-aligning  plunger 
line  switch  and  master  switch  circuit 
and  of  the  wiring  arrangement  be- 
tween trunks  and  line  switches. 
Though  only  three  line  switches  are 
shown  in  this  latter  arrangement, 
there  may  be  from  25  to  100  such 
switches  in  one  group.  Figures  7 
and  8  show  views  of  the  line  and 
master  switches. 

When  the  handset  is  lifted  at  the     «       0      **•    A      «    -j.  i.     *        •  ^  -,      -,«!,. 

FIG,   8.     Master   S-witch   Associated    with   Self-i,_0 „ 

Plunger  Line  Switch  (Courtesy  Automatic  Electric  CoJ 


telephone,  the  plunger  is  thrust  into 

the  line  switch  bank  by  operation  of 

the  A  and  B  relays,  closing  the  line  and  trunk  spring  contacts  and  extending  the  line 

through  to  the  selector.     The  operation  of  the  selector  relays  maintains  the  B  relay  and 


MECHANICAL  SYSTEMS  AND  OPERATION 


17-11 


its  plunger  operated  until  the  connection  is  released,  the  relays  then  returning  to  normal. 
Operation  of  the  plunger  starts  operation  of  the  master  switch  circuit,  resulting  in  the 
moving  of  all  idle  line  switch  plungers  opposite  the  next  idle  trunk.  The  self-aligning 
feature  of  the  plunger  reseats  it  on  the  master  switch  guide  bar  as  soon  as  it  is  released. 
The  connector  bank  terminal  associated  with  the  line  is  also  made  busy  by  the  selector 
placing  ground  on  the  control  lead. 

The  rotary  line  switch  is  a  single-motion  device,  which  may  be  associated  with  a  tele- 
phone line  for  the  purpose  of  extending  the  line  to  any  one  of  a  number  of  idle  trunks.  This 
switch  has  a  shaft  carrying  wipers,  which  slide  over  bank  contacts  (arranged  in  a  half 
circle)  to  which  trunks  to  the  numerical  switches  are  connected.  When  the  line  and  cut-- 
off relays  are  energized  by  lifting  the  handset,  the  switch's  driving  magnet  operates  its 

TO  CONNECTOR  1 

SI® 

TOO  LINES 


TO  .TRUNK  - 
SWITCH     (g) 


FIG.  9.     Schematic  Diagram  of  the  200  Line  Finder  Circuit  (Courtesy  Bell  System) 

armature  against  the  action  of  a  fiat  spring.  At  the  end  of  the  armature  stroke  the  magnet 
circuit  is  opened,  the  spring  forces  the  armature  back,  and  the  wipers  are  carried  forward 
one  step  into  the  bank.  This  action  continues  until  the  wipers  reach  idle  trunk  contacts, 
where  they  remain  until  the  nest  call  is  originated,  when  the  wipers  are  moved  to  the  next 
idle  trunk.  It  is  usually  necessary  to  extend  connections  by  means  of  wipers  and  bank 
contacts  for  several  conductors  which  perform  signal  or  control  functions  in  addition  to 
extension  of  the  transmission  path.  Thus,  each  position  of  the  wipers  in  the  bank  has 
from  three  to  six  or  more  bank  contacts,  which  are  simultaneously  contacted  by  separate 
wipers  when  the  selection  is  made.  This  type  of  switch  is  seldom  used  as  a  subscriber 
line  switch  since  the  plunger  type  is  cheaper,  but  both  operate  satisfactorily. 

Secondary  line  switches,  rotary  or  plunger  type,  are  now  seldom  used  but  were  designed 
for  larger  step-by-step  offices  to  combine  the  traffic  from  a  number  of  primary  groups  of 
subscriber  line  switches  and  direct  it  to  a  relatively  large  common  group  of  numerical 
switches  capable  of  handling  the  combined  traffic.  Small  trunk  groups  are  less  efficient 
than  large  trunk  groups,  and  by  employing  secondary  line  switches  the  number  of  se- 
lectors required  for  the  main  trunk  group  need  be  only  large  enough  to  accommodate  the 
peak  load  of  the  combined  group  instead  of  each  small  group  requiring  enough  selectors 
to  accommodate  its  own  peak  load,  which  usually  will  not  occur  at  the  same  time  as  the 
peak  loads  of  the  other  small  groups. 

The  line  finder  switch  seeks  out  the  calling  line  from  a  group  of  subscriber  lines  connected 
to  bank  contacts  and  connects  it  to  a  trunk  terminating  in  the  first  numerical  switch  of  the 


17-12 


TELEPHONY 


switch,  train.  The  line  finder  switch,  permits  the  use  of  a  simple,  economical  subscriber 
line  circuit  composed  principally  of  a  line  and  a  cutoff  relay  comparable  to  those  in  com- 
mon-battery manual  systems.  These  relays  function  to  mark  the  bank  position  of  the  call- 
ing line  and  to  cause  the  allotted  line  finder  to  hunt  the  calling  line. 

As  soon  as  the  bank  position  of  a  calling  line  is  marked  by  operation  of  the  line  relays, 
a  relay  of  a  common  relay  group  (associated  with  each  group  of  line  finders)  causes  the 
proper  line  finder  to  hunt,  first  vertically  and  then  horizontally,  until  the  calling  line  is 
located,  whereupon  the  finder  connects  its  permanently  associated  first  selector  (or  con- 

nector) to  the  calling  line  through  the  finder  wipers  and 
the  bank  contacts.  The  cutoff  relay  of  the  line  circuit  then 
operates  to  clear  the  line  of  unnecessary  attachments  and 
to  free  the  common  group  of  relays. 

A  number  of  line  finders  are  grouped  under  the  control 
of  a  single  distributor,  which  preselects  or  allots  the  next 
line  finder  to  be  used  in  the  next  call  as  soon  as  the  com- 
mon relay  group  is  released  from  the  preceding  call. 

Line  finder  switches  are  of  the  two-motion  (vertical  and 
horizontal)  type  and  usually  have  capacities  for  50,  100, 
or  200  lines.  In  some  small  exchanges,  line  finders  may 
be  of  the  rotary  (single-motion)  type  with  capacities  of  25 
or  50  lines. 

Figure  9  shows  diagrammatically  the  switching  arrange- 
ment between  subscriber  lines  and  trunks,  and  Pig.  10  a 
view  of  a  line  finder  switch,,  for  200-line  capacity.  This 
switch  has  a  group  of  relays  mounted  on  a  base,  upon 
which  is  also  mounted  a  frame  supporting  a  shaft  with 
ratchet  mechanism  for  raising  and  rotating  the  shaft.  The 
lower  part  of  the  shaft  carries  four  sets  of  Wipers  (one 
single-conductor  and  three  two-conductor),  termed  the 
vertical,  control  (upper  bank),  upper  line  (middle  bank),  and 
lower  line  (lower  bank).  The  vertical  and  rotary  stepping 
magnets  and  the  release  magnet  (which  permits  the  shaft 
to  return  to  normal  when  the  connection  is  released)  are 
mounted  within  the  switch  frame. 

A  vertical  interrupter  (pulsing)  circuit  causes  the  vertical 
magnet  (by  its  armature  and  pawl  engaging  the  "vertical 
hub"  ratchet)  to  elevate  the  shaft  step  by  step  to  the 
marked  level.  A  rotary  interrupter  circuit  then  causes  the 
^  m^f  <ft  «•  -mature  and  pawl  engaging  the 
rotary  hub  ratchet)  to  rotate  the  shaft  step  by  step, 
until  the  marked  control  bank  contact  is  engaged  by  the 
control  wiper. 

These  motions  cause  the  control,  upper  line,  and  lower  line  wipers  to  engage  the  cor- 
responding contacts  of  their  semicylindrical  banks,  each  of  which  has  100  sets  of  contacts 
(10  levels  and  10  sets  per  level).  To  the  right  of  these  banks  is  the  vertical  bank  or  com- 
mutator, comprising  a  single  row  of  contacts,  over  which  the  vertical  wiper  moves  until 
the  marked  level  of  the  calling  line  is  reached. 

The  release  of  the  shaft  is  effected  by  the  operation  of  the  release  magnet,  which  dis- 
engages the  vertical,  rotary,  and  stationary  dogs  from  their  ratchets,  permitting  the  shaft 
to  return  to  normal  under  spring  and  gravity  action. 

The  impulse  repeater  is  used  in  interoffice  trunks  in  step-by-step  exchange  areas  having 
more  than  one  office.  This  repeater  is  required  in  the  outgoing  end  of  each  trunk  and 
functions  (1)  to  make  it  unnecessary  to  provide  a  third  (control)  wire  in  each  trunk,  (2) 
to  provide  talking  current  to  the  calling  subscriber,  (3)  to  reverse  battery  to  the  calling 
subscriber  when  the  called  subscriber  answers,  and  (4)  to  repeat  dial  pulses  over  the  inter- 
office trunk  so  that  the  impulse  circuit  will  not  include  both  the  subscriber's  line  and  the 
interoffice  trunk. 

The  two  types  of  impulse  repeaters  are  one-way  and  two-way.  The  first  type  is  used' 
at  one  end  of  one-way  interoffice  trunks;  the  second,  at  both  ends  of  two-way  interoffice 
trunks. 

A  diagram  of  the  one-way  impulse  repeater  circuit  is  shown  in  Fig.  11. 

When  the  repeater  is  seized  by  the  preceding  switch,  the  A  and  B  relays  operate,  and 

the  B  relay  connects  ground  to  the  control  lead  C,  to  protect  and  hold  the  preceding 

switches  in  the  train  and  avoid  seizure  by  other  switches.     R,elay  B  also  operates  relays 

A-l  and  B-l  in  the  incoming  selector  at  the  distant  office,  establishing  an  impulse  loop. 


(Courtesy 


Automatic 
Co.) 


Electric 


MECHANICAL  SYSTEMS  AND   OPERATION 


17-13 


over  the  trunk.  Relay  A,  responding  to  impulses  from  the  calling  subscriber  dial,  in- 
terrupts the  impulse  loop,  according  to  the  impulses  received,  thereby  repeating  these 
impulses  over  the  loop. 

When  the  called  subscriber  answers,  operation  of  the  back-bridge  relay  of  the  connector 
at  the  called  office  reverses  the  polarity  of  the  current  through  the  holding  bridge  (relay 
F)  of  the  repeater  at  the  calling  office,  causing  relay  F  to  operate.  The  operation  of  relay 
F  causes  relay  D  of  the  repeater  to  operate,  which  reverses  the  polarity  of  the  curreni  now 
to  the  calling  telephone,  for  the  purpose  of  operating  coin-collectors  or  message  registers 
or  of  providing  supervision  of  manual  calls. 

When  the  handset  at  the  calling  telephone  is  placed  on  its  support,  the  relays  release 
and  the  train  of  switches  is  restored  to  normal. 

The  selector  is  a  switching  device  which  became  necessary  for  offices  of  over  100  lines. 
In  a  1000-termmal  system,  only  a  single  rank  of  selectors  (first)  is  required,  the  first  digit 
dialed  operating  the  selector  switch  to  connect  to  the  desired  hundred  group  of  connectors. 


LOCAL  OFFICE 


DSTANT  QPFtCE 


NOTE:    RELAY  »F"  MUST   NiOT  PULL  UP   UtsTTtL 

BATTERY  S  REVERSED  OVER  THE  TRUNK 


THJUW   PPi  AY 
COt!iT*CTS 


FIG.  11.     One-way  Impulse  Repeater  Circuit  —  Strowger  System  (Courtesy  Automatic  Electric  Co.') 

In  a  10,000-terminal  system,  two  ranks  of  selectors  (first  and  second)  are  required.  The 
first  digit  dialed  operates  the  first  selector  to  select  the  thousand  group  of  trunks  which 
terminate  in  second  selectors,  and  the  second  digit  dialed  operates  the  second  selector  to 
select  the  desired  hundred  group  of  connectors.  Thus,  the  selector  is  a  numerical  type  of 
group-selecting,  trunk-hunting  two-motion  switch,  which  requires  but  one  digit  for  its 
operation. 

Figure  12  shows  a  schematic  diagram  of  the  selector  circuit.  The  selector  has  a  group 
of  control  relays  mounted  on  a  base,  which  also  supports  a  shaft  and  ratchet  mechanism 
assembly  for  raising  and  rotating  the  shaft.  The  lower  part  of  the  shaft  carries  two  sets 
of  wipers,  control  (upper)  and  line  (lower).  The  vertical  and  rotary  (stepping)  magnets 
and  the  release^magnet  are  mounted  within  the  switch  frame.  The  bank  contacts  are  in 
two  groups  of  100  sets  of  contacts  each  (10  levels  and  10  sets  per  level)  . 

When  the  selector  is  seized,  it  functions  to  hold  all  preceding  switches  in  the  train  oper- 
ated and  guarded  until  the  holding  circuit  is  extended.  It  sends  back  dial  tone  to  the 
calling  subscriber  if  it  is  a  first  selector.  It  elevates  the  shaft  and  wipers  in  response  to 
dial  pulses  and  rotates  them  automatically  to  connect  with  an  idle  trunk  in  the  selected 
bank  level.  It  provides  a  busy  signal  to  the  calling  subscriber  when  all  trunks  in  the  de- 
sired group  are  busy.  The  selector  is  returned  to  normal,  when  the  calling  subscriber 
places  his  handset  on  its  support,  by  functioning  of  the  control  circuit  and  the  release 
magnet  of  the  selector. 

The  connector  is  a  two-motion  switch,  similar  to  the  selector,  and,  regardless  of  the  size 
of  the  office,  it  is  always  employed  as  the  final  unit  of  step-by-step  switch  trains. 

This  switch  operates  in  response  to  the  last  two  digits  dialed  of  the  called  number 
(directory  listing).  The  first  of  these  two  digits  is  the  "tens"  and  the  last  one  the  "units" 
digit.  The  only  exceptions  to  this  general  principle  are  the  200-line  connector  and  the 
frequency  or  code-selecting,  party-line  connectors,  where  a  digit  preceding  or  succeeding 
the  "tens"  and  "units"  digits  is  dialed  for  line  group  or  ringing  selection. 

Figure  13  shows  a  schematic  diagram  of  the  connector  circuit.  The  connector  has  a 
group  of  control  relays  mounted  on  a  base,  which  also  supports  the  shaft  and  ratchet 


17-14 


TELEPHONY 


SUPERVISORY  RELAY 

&  NEGATIVE  'BATTERY 


OFF -NORMAL  CONTACTS 
CLOSE  WITH  FIRST  VERTICAL 
STEP  JUST  BEFORE  DOUBLE 
DOG  FALLS    IN 


CAM  SPRINGS  (SWITCHED 
ON  11TH  ROTARY  STEP  BY 
SWITCH  SHAFT  CAM) 


1000W  NON 
INDUCTIVE 


SUPERVISORY  RELAY   &, 
NEGATIVE    BATTERY 


DIAL  TONE  AND  GROUND 


FIG.  12.     Selector  Circuit — Strowger  System  (Courtesy  Automatic  Electric  Co.) 


RING-BACK  TONE 


1  CAPACITOR 
PER  CONNECTOR  GROUP 


FIG.  13.     Connector  Circuit — Strowger  System  (Courtesy  Automatic  Electric  Co.) 


MECHANICAL  SYSTEMS  AND  OPERATION  17-15 

mechanism  assembly  for  raising  and  rotating  the  shaft,  step  by  step.  The  lower  part  of 
the  shaft  carries  two  sets  of  wipers,  the  control  (upper)  and  line  (lower!,  which  engage 
respective  semicircular  banks  of  contacts  of  100  sets  each  (10  levels  and  10  sets  per  level*. 
The  vertical  and  rotary  stepping  magnets  and  the  release  magnet  are  mounted  within  the 
switch  frame.  One  subscriber  line  is  connected  to  each  set  of  lower  line  bank  contacts. 

The  pulses  of  the  first  digit  received  (except  as  mentioned  above*  siep  the  shaft  and 
wipers  vertically  as  many  levels  as  there  are  pulses.  The  pulses  of  the  last  digit  received 
step  the  shaft  and  wipers  horizontally,  in  accordance  with  the  number  of  pulse*  reeled. 
This  desired  position  of  the  shaft  assembly  is  held  by  two  movable  detents,  termed  the 
"double-dog,"  and  a  stationary  dog. 

^  The  release  of  the  shaft  is  under  the  control  of  the  release  magnet,  which,  in  operating, 
disengages  the  dogs,  allowing  the  shaft  assembly  to  return  to  normal. 

The  director  system  was  originated  and  developed  by  Automatic  Electric  Co.  for  large 
and  complex  trim  king  networks  of  the  Strowger  system.  Though  not  used  in  the  United 
States  to  date,  it  has  extensive  application  in  Great  Britain,  particularly  in  the  London 
metropolitan  area.  It  is  being  considered  in  the  Los  Angeles  area  for  certain  S  X  S  offices 
to  meet  extended  service  and  automatic  toll  ticketing  problems.  The  director  is  ex- 
pected to  play  an  important  part  in  nationwide  toll  dialing,  requiring  register-sender  equip- 
ment. 

The^director,  itself,  consists  of  standard  A.  E.  Co.  relays  and  switches,  which  store  the 
subscriber  dial  pulses  and  perform  various  other  functions.  A  director  for  simple  functions 
occupies  the  space  of  two  regular  switches,  but,  being  very  flexible  in  design,  it  may  be  of 
various  sizes  for  specific  needs.  Wherever  used,  this  unit  usually  effects  savings  in  se- 
lectors, repeaters,  and  floor  space.  It  can  be  added  to  existing  S  X  S  equipments  as  de- 
sired. 

In  operation,  an  idle  director  is  selected  by  a  director  finder  and  attached  to  a  line  finder- 
first  selector  trunk  as  soon  as  the  line  finder  seizes  the  calling  line.  The  director  then 
functions  as  an  "electrical  brain"  to: 

1.  Record  the  number  (pulses)  dialed  by  the  subscriber, 

2.  Analyze  the  office  code  digits  received  and  immediately  determine  the  best  routing 
for  the  call. 

3.  Substitute,  if  necessary,  other  routing  digits,  which  may  differ  entirely  from  the  re- 
ceived digits.     This  is  known  as  translation,  which  permits  automatically  selected  alter- 
native routings  with  resulting  trunk  savings. 

4.  Send  out  pulses,  corresponding  to  the  routing  code,  which  operate  switches,  as  though 
operated  by  the  subscriber  dial. 

5.  Store  the  remaining  digits,  dialed  by  the  subscriber,  ustially  without  translation,  and 
send  out  corresponding  pulses  just  after  the  routing  code  is  sent. 

6.  Detach  itself  from  the  connection  upon  completion  of  operation  5  and  await  the  nest 
call. 

The  toll  switch  train,  consisting  of  a  toll  transmission  selector,  a  toll  intermediate  selector, 
and  a  combination  toll  and  local  connector,  is  designed  to  complete  toll  calls  directly  from 
the  toll  board  to  the  subscriber.  The  toll  transmission  selector  provides  increased  talking 
current  to  the  called  subscriber  telephone  through  repeating  coil  windings  and  has  in- 
creased capacity  in  the  talking  circuit.  It  also  provides  for  complete  supervision  of  the 
connection  at  the  toll  board,  and  it  repeats  the  dial  pulses  from  the  toll  operator  dial 
through  to  the  toll  intermediate  selector  and  the  toll  connector.  It  has  a  400-point  bank, 
required  for  additional  functions.  The  toU  intermediate  selector  is  similar  to  the  regtilar 
selector  except  that  it  also  has  a  400-point  bank.  The  combination  connector  is  not  used 
for  local  calls  unless  all  the  regular  connectors  are  in  use. 

Main  distributing  frames  (MDF)  and  intermediate  distributing  frames  (IDF)  provide  a 
means  for  properly  terminating  the  outside  cables,  which  carry  subscriber  liaes  and  various 
types  of  trunks  and  toll  circuits,  and  equipment  within  the  office,  and  cross-coonectiiig 
these  various  circuits  and  equipments  as  required. 

Multipling  of  trunks  and  of  switching  equipment  is  so  arranged  as  to  provide  maximum 
access  to  switches  from  subscriber  lines  and  from  switches  to  lines,  and  efficient  operation 
of  trunking  and  other  equipment. 

Figure  14  shows  a  schematic  diagram  of  a  trunk  layout  for  a  100,000-line  multiofBce 
exchange.  It  will  be  noted  that  secondary  line  switches  are  employed  between  the  primary 
line  switches  and  first  selectors  and  between  the  first  selectors  and  impulse  repeaters,  the 
object  being  to  concentrate  the  traffic  loads  in  these  sections  and  reduce  to  a  minimum  the 
number  of  first  selectors  and  repeaters  required.  All  manual  board  services  are  centralized 
at  one  office  for  efficient  operation.  A  special  switch  train  is  provided  to  reduce  wrong 
numbers  which  may  result  from  careless  removal  of  a  handset  or  accidental  jiggling  of  the 
cradle  plunger  switch.  This  equipment  is  not  always  warranted. 


17-16 


TELEPHONY 


The  U  suboffice  represents  a  telephone  center  of  light  load,  located  rather  remotely 
from  the  other  offices.  Traffic  does  not  justify  a  separate  group  of  trunks  to  each  of  the 
other  offices,  so  that  the  office  is  made  tributary  to  office  U.  To  reach  a  subscriber  in  the 


office,  C/0  is  dialed,  the  first  incoming  switch  in  the  U  suboffice  being  a  third  selector. 
Outgoing  calls  from  this  office  pass  through  the  main  U  office. 

Multioffice  tarunking  and  other  switching  in  the  Strowger  system  is  accomplished,  using 
in  combinations  the  types  of  step-by-step  switches  and  repeaters  described  above  and,  in 


MECHANICAL  SYSTEMS  AND  OPERATION  17-17 

addition,  many  other  auxiliary  circuits  and  devices  which  limited  space  will  not  permit  dis- 
cussing here. 

THE  PANEL  DIAL  SYSTEM  is  a  radical  change,  both  in  mechanical  and  electrical 
characteristics,  from  the  Stronger  system  and  is  employed  generally  in  the  largest  metro- 
politan areas.  Great  flexibility  of  operation  is  permitted  by  this  system  as  trunk  groups 
and  their  sizes  are  for  all  practical  purposes  arbitrary.  The  time~of  establishing  a  con- 
nection is  not  dependent  on  the  step-by-step  dialing  process  of  the  subscriber,  since,  after 
all  the  dialing  pulses  are  received  from  the  subscriber  by  an  intricate  mechanism,  the  con- 
nection is  rapidly  routed  to  the  called  line,  under  control  of  this  same  mechanism,  with 
great  speed.  The  equipment  as  a  whole  is  necessarily  complex. 

The  principal  mechanisms  in  the  system  are: 

1.  The  panel-type  selector  with  its  banks  of  line  or  trunk  terminals,  which  are  selected 
by  vertically  moving  brushes  mounted  on  bra^s  rods. 

2.  The  sequence  switch,  which  has  a  number  of  insulating  disks  mounted  on  a  shaft. 
Each  disk  has  a  metal  stamping  on  each  side,  and  the  entire  assembly  of  disks  turns  a 
step  at  a  time,  as  directed  by  the  control  circuit,  with  two  brushes  bearing  on  each  stamp- 
ing.    By  this  means  a  large  number  of  circuit  arrangements  may  be  made  aa  required  to 
establish  the  talking  connection  at  the  proper  time,  as  determined  by  the  control  circuit. 

3.  The  decoder  sender  is  the  "brains"  of  the  system.     It  registers  and  stores  the  dial 
pulses  from  the  subscriber  dial  by  means  of  a  dial  pulse  register  circuit;  it  decodes  the 
numerical  digits  dialed  into  a  non-numerical  selection  scheme  by  chooang  the  proper 
route  relay.     The  route  relay  controls  the  selection  of  an  interoffice  trunk.     The  sender 
then  takes  over  control,  sending  out  pulses  to  actuate  a  train  of  selectors  which  establish 
connection  with  the  called  line.     The  decoder  sender  is  then  released  for  other  calls. 

The  panel-type  selector,  from  which  the  system  derives  its  name,  performs  the  same 
function  in  this  system  that  the  line  finder  and  selector  switches  perform  in  the  Strowger 
system.  The  panel-type  selector  frame  consists  of  several  panel  multiple  banks  of  sub- 
scriber line  or  trunk  terminals,  over  which  the  selector  brushes  slick  vertically,  and  mech- 
anism for  moving  and  controlling  the  selector  motion,  as  shown  in  Fig.  15. 

The  panel  multiple  bank  consists  of  horizontally  projecting  terminals,  arranged  in  "ver- 
tically positioned,  rectangular  panels  or  banks,  as  shown  in  Fig.  16.  Each  terminal  in 
the  panel  consists  of  a  fiat  brass  strip  extending  horizontally  through  the  panel  and  having 
30  projections  on  each  side  of  the  panel  and  also  a  soldering  lug  at  each  end  of  the  atrip 
for  wiring.  A  set  of  three  of  these  strips,  mounticd  one  above  the  other  and  insulated 
from  each  other  with  impregnated  paper,  constitute®  the  tip,  ring,  and  sleeve  terminals 
of  one  line.  Thus,  each  line  appears  horizontally  across  the  face  of  the  panel  30  times, 
or  60  times  for  both  faces.  The  number  of  lines  or  trunks  provided  in  the  banks  varies 
in  accordance  with  requirements,  present  practice  with  respect  to  subscriber  lines  being 
to  provide  40  lines  per  panel  and  10  panels  per  frame. 

A  selector  is  placed  opposite  each  three  vertical  rows  of  line  terminals;  it  consists  of  a 
hollow  vertical  brass  tube  on  which  are  fastened  ten  sets  of  spring  brushes,  one  set  of 
brushes  for  each  terminal  bank.  Each  brush  has  three  contacts,  normally  held  apart  by 
an  insulator  so  as  not  to  touch  the  lugs  or  terminals  on  the  terminal  panels.  Each  brush 
contact  is  connected  in  multiple  with  the  corresponding  cant  act  of  the  other  nine  sets  of 
brushes  on  the  same  brass  tube  or  selector,  so  that  any  selector  may  reach  any  one  of  the 
400  lines  in  the  frame  by  tripping  the  proper  brush,  and  the  total  selector  movement  will 
not  exceed  one  of  the  ten  banks  of  terminals.  In  practice,  each  bank  of  40  lines  is  divided 
vertically  into  two  identical  banks  of  the  same  lines,  but  with  the  line  numbering  in  reverse 
order  (bottom  to  top  and  top  to  bottom  of  the  bank) .  This  arrangeji&eBt  of  line  number- 
ing still  further  reduces  the  travel  of  the  selector  brushes  to  reach  a  particular  line  and 
reduces  operating  time  of  the  selectors.  A  trip  rod  is  provided  with  trip  levers  for  each 
bank,  so  that  the  brush  which  is  selected  to  make  contact  with  the  calling  Ene  is  tripped  as 
it  starts  upward  in  the  calling  line  bank,  the  insulator  between  the  brush  contacts  is  with- 
drawn, and  these  contacts  make  the  connection  with  the  calling  line  terminals  when  they 
are  reached. 

The  multiple  wiring  between  the  brushes  is  contained  inside  the  hollow  bras©  tube  of 
the  selector  and  terminates  in  another  set  of  brushes  at  the  top  of  the  frame,  which  slide 
on  bars  in  the  commutator  panel  and  control  the  selector  movement. 

The  control  mechanism  of  the  selector  is  located  at  the  bottom  of  the  frame.  There 
are  two  horizontal  cork-covered  rolls,  extending  the  width  of  the  frame  and  driven  in  op- 
posite directions  by  suitable  gears  and  a  motor.  Attached  to  the  lower  end  of  each  se- 
lector tube  is  a  fiat  bronze  strip  or  rack,  which  is  close  to,  but,  in  the  idle  position,  not 
touching,  the  continuously  revolving  cork  rolls.  An  eleetromagneticallr  operated  clutch 
is  mounted  on  the  selector  frame  in  front  of  each  rack.  Energisation  of  either  the  up  or 
down  magnet  presses  the  rack  against  the  up  or  down  cork  roll,  causing  the  selector  to 


17-18 


TELEPHONY 


move  up  or  down  as  desired.  A  pawl  (not  shown)  engages  in  the  horizontal  slits  of  the 
rack  and  prevents  the  selector  from  slipping  down  after  it  has  been  elevated  to^the  proper 
level.  The  selector  returns  to  its  normal  position  when  released  by  the  trip  magnet. 
Figure  16  shows  the  control  mechanism. 


Rear  of  seq.  sw.  & 

relays  on  other 

ide  of  frame 


Drive  Motor 


FIG.  15.     General  View  of  Panel  Frame — Panel  Dial  System  (Courtesy  Bell  System) 

Control  of  the  up-and-down  movement  of  the  selector  is  obtained  either  from  impulses 
coming  from  the*  sender  or  from  connections  made  on  the  commutator  at  the  top  of  the 
frame.  Panel  selector  equipment  may  vary  in  construction  details  and  wiring,  depending 
on  its  assigned  function  in  the  system. 


MECHANICAL  .SYSTEMS  AND  OPERATION 


17-19 


Soldering  lug 


I—L. — I — WJJ__U — U — U — IJJ-H 

X^^^     Soldering  lugs 
X  Ad  justing  screw 


Aluminum  comb 
Panel  Multiple  Bank 


Sddenng  lugs 


Selector- 


Trip 
Magnet 


-Rack 


'Cork  Ro»s 


Up  Drive 
Magnet 


BoIJer  t 
Selector  Control  Mechanism 


Driven  Disc 

-  Spring  RoJter 


Drive 
Magnet 


Cam  Designations 

Stotfor 
Brush  Adjustment 


index 
Index  Wheel 


Brush  Ctamp 


"A"  CAM 


TYPICAL  CAM  BRUSH  ASSEMBLY 

Sequence  Switch 
FIG.  16.     Panel  Frame  Units,  Showing  Details—Panel  Dial  System  (Courtesy  Bell  System) 


17-20 


TELEPHONY. 


TO  LINE  CIRCUIT 
I 


The  sequence  switch.  Is  vital  to  panel  operation.  The  control  circuits  for  panel-type 
selectors  are  necessarily  complex  and  must  be  set  up  in  sequence  to  cause  the  various 
operations  for  establishing  a  connection  to  take  place  at  exactly  the  right  time.  This 
switch  (Fig.  16)  has  24  disks  or  cams  mounted  permanently  on  a  shaft,  and  each  disk  con- 
sists of  insulation  with  a  specially  shaped  metal  stamping  attached  to  each  side  of  the  disk. 
Two  brushes  bear  on  each  side  of  each  disk.  The  metal  stampings  are  of  different  shapes, 
so  that  as  the  disks  turn  each  brush  may  rest  on  the  stamping  or  on  the  insulation,  in  ac- 
cordance with  the  position  of  the  switch.  This  provides  a  means  of  establishing  con- 
nections between  brushes  or  opening  and  closing  circuits  in  various  combinations,  as 
desired.  A  fluted  metal  disk,  on  which  a  spring  roller  rides,  is  mounted  at  the  end  of  the 
shaft,  so  that  the  shaft  may  be  revolved  and  held  hi  any  one  of  18  positions.  An  electro- 
magnet, when  energized,  pulls  this  disk  against  the  edge  of  another  continuously  revolving 

disk  and  thus  causes  the  shaft  to  turn  one 
step  at  a  time  for  each  energization  of  the 
electromagnet. 

One  sequence  switch  is  associated  with 
each  selector  and  is  located  on  the  right  of 
the  terminal  bank,  as  shown  in  Fig.  15. 

The  decoder  sender  records  the  pulses 
dialed  by  the  subscriber,  translates  and  de- 
codes the  first  group  of  pulses  (office  code), 
resulting  in  selection  of  the  proper  central 
FIG.  17.  Interconnection  of  Decoder  Sender  office  trunk  or  trunks,  and  finally  controls 
Units-Panel  Dial  System  (Courtesy  Bell  Sys-  the  brush  selection  and  functioning  of  the 

various  selectors  and  the  sequence  switch. 

The  major  units  consist  of  a  sender,  decoder  connector,  and  decoder,  as  shown  in 
Fig.  17. 

The  dial  pulse  register  circuit  of  the  sender  consists  of  a  group  of  relays  so  connected 
as  to  record  the  dial  pulses  and  subsequently  control  the  operation  of  the  decoder  and 
selectors  in  establishing  the  connection.  Figure  18  shows  the  portion  of  the  dial  pulse 
register  circuit  which  registers  the  first  three  series  of  pulses  comprising  the  office  code. 
The  connection  of  the  sender  to  the  subscriber's  line  operates  the  AC  relay,  so  that  the 
P  group  of  relays  is  connected  to  the  A  group.  The  P  group  counts  the  pulses  as  received 
from  the  subscriber  dial.  The  release  and  reoperation  of  the  L  relay,  under  control  of  the 
dial,  operate  the  P  relays.  The  A  recording  relays  are  operated  by  the  action  of  the  P 
relays  in  such  manner  that  the  sum  of  their  numbers  is  equal  to  the  figure  dialed.  Thus, 
if  the  first  letter  of  the  office  name  is  S,  causing  seven  pulses  to  be  sent,  relays  A-2  and 
A-5  will  operate.  When  the  first  figure  has  been  recorded,  the  RA-l  relay  supplies 
ground  to  the  A  recording  relays,  locking  them,  and  releases  the  P  relays.  The  P  relays 
are  then  connected  to  the  B  recording  relays,  which  are  ready  to  record  the  next  figure 
dialed.  This  operation  is  similar  for  each  of  the  remaining  figures  dialed  by  the  sub- 
scriber. 

There  are  eight  groups  of  recording  relays,  three  for  the  office  code  and  five  for  numerical 
digits.  The  last  five  groups  are  not  shown  in  Fig.  18,  but  they  are  similar  to  those  shown. 

When  three  digits  have  been  dialed,  relay  CL  operates  (relay  Az  operates  in  place  of 
relay  CL  if  the  first  figure  dialed  is  zero).  The  operation  of  either  of  these  relays  indicates 
that  the  sender  is  ready  for  translation.  The  sender  is  now  automatically  connected  to  a 
decoder,  through  the  decoder  connector,  by  connection  of  the  A,  B,  and  C  groups  of  leads 
shown  in  Fig.  18  to  the  corresponding  groups  of  leads  shown  in  Fig.  19. 

The  decoder  register  relay  groups,  A,  B,  and  C,  shown  in  Fig.  19,  then  register  the 
hundreds,  tens,  and  units  digits  of  the  office  code,  stored  in  the  sender.  The  setting  of  the 
A  register  relays  causes  one  of  the  multicontact  relays,  H-2  to  l?-9,  to  operate;  the 
setting  of  the  B  relays  causes  operation  of  the  proper  T  relay,  and  the  setting  of  the  C 
relays  grounds  one  of  the  10  leads  shown  at  the  right  in  Fig.  19. 

Each  jET  relay  has  100  armatures  and  contacts  as  indicated  in  Fig.  19.  The  800  con- 
tacts are  connected  to  terminal  strip  punchings,  called  code  points.  The  operation  of  an 
H  relay  connects  the  proper  hundred  code  points  to  the  hundred  contacts  of  the  T  relays 
(ten  to  each  relay).  The  operation  of  the  proper  T  relay  connects  ten  code  points  to  the 
ten  leads  from  the  C  relays,  and  the  setting  of  the  C  relays  connects  ground  to  one  code 
point. 

The  code  points  are  cross-connected  to  route  relays  (Fig.  19),  one  of  which  is  provided 
for  each  possible  path,  which  a  call  may  take.  Since  a  particular  route  relay  may  be  cross- 
connected  to  any  code  point,  so  that  the  selection  of  a  particular  route  relay  is  numerical, 
the  selection  operation  from  this  point  on  need  not  be  on  the  decimal  basis  as  indicated 
above. 


MECHANICAL  SYSTEMS  AND  OPERATION 


17-21 


fr  f*  •  !f  y  1S  S°  desi^ed  that  it  requires  certain  operations  of  the  district 
th ™h  selectorS_mdependently  of  the  code  point  to  which  it  is  connected,  and  thus  of 
the  number  dialed  in  securing  it.  If  the  position  of  a  group  of  trunks  i*  to  be  changed  on 
the  selector  the  route  relay  must  be  changed  accordingly,  but  no  further  change  in  the 
sj  stem  need  be  made.  Only  one  route  relay  is  required  in  each  sender  for  each  outgoing 


trunk  group  from  the  district  and  office  multiple,  that  is,  one  for  each  dialing  combination 
used  in  the  particular  exchange.  The  route  relay,  through  a  cotnbifiatioa  of  registering 
relays,  determines  the  brushes  to  be  tripped  and  the  movements  of  the  selectors  which 
choose  the  proper  office.  Once  the  proper  route  relay  is  selected,  the  remaining  operation 
of  obtaining  the  proper  office  is  straightforward.  All  vacant  code  points  are  strapped  to- 
gether and  connected  to  a  single  route  relay,  which  will  cause  the  sender  to  complete  the 
call  to  an  operator,  who  will  explain  to  the  subscriber  that  an  incorrect  number  has  been 
dialed. 


17-22 


TELEPHONY 


MECHANICAL  SYSTEMS  AND  OPERATION 


17-23 


When  the  decoder  has  completed  translation,  or  if  translation  cannot  be  completed 
owing  to  trouble,  the  decoder  signals  the  sender,  which  releases  the  decider.  Thus,  the 
decoder  is  needed  only  during  the  time  the  office  trunk  is  being  chosen,  but  the  sender  is 
needed  in  the  connection  until  the  called  subscriber  line  is  connected.  More  senders  will 
be  required  than  decoders,  the  usual  ratio  being  about  400  senders  to  not  over  10  decoders, 
although  this  ratio  varies  with  the  traffic  load. 

IN  PANEL  DIAL  OPERATION  each  subscriber  line  terminates  on  a  line  finder  frame 
for  handling  outgoing  calls,  and  a  final  frame  for  handling  incoming  calls,  as  shown  in  Fig, 

20.  When  a  subscriber  lifts  his  handset  to  call,  direct  current  from  the  central-office 
battery  flows  through  his  line  and  operates  the  associated  line  relay,  at  the  same  time 
making  his  line  busy  to  any  selector  on  the  final  frame.     The  operation  of  the  line  relay 
causes  an  idle  line  finder  selector  associated  with  the  bank  of  terminals  in  which  his  line 
appears  to  move  upward  until  it  finds  the  calling  line;  this  relay  also  causes  the  trip  rod  in 
that  bank  to  trip  the  proper  brush  of  the  moving  selector,  after  which  the  trip  rod  is  im- 
mediately reset. 

Simultaneously  with  the  motion  of  the  selector,  the  panel  link  circuit  is  selecting  the 
district  selector  associated  with  the  line  finder  selector,  and  also  an  idle  sender.  The 
panel  link  frame  is  also  similar  to  the  panel  type  selector  frame,  but  with  different  details 
to  suit  its  functions.  The  frame  is  divided  into  two  equal  parts  vertically  with  the  sender 
selectors  on  the  left  and  the  district  selectors  on  the  right.  Each  district  selector  is  wired 
directly  to  its  corresponding  sender  selector. 

In  practice,  the  district  selector  when  released  from  a  call  selects  the  next  idle  line  finder, 
while  the  sender  selector  remains  where  it  is  unless  near  the  top  of  the  bank,  in  which  case 
it  returns  to  normal  position. 

When  the  line  finder  reaches  the  calling  line  and  the  sender  selector  has  found  an  idle 
sender,  dial  tone  is  sent  back  to  the  calling  subscriber. 

The  action  of  the  sender  is  described  in  detail.  The  dial  pulses  are  stored  in  the  dial 
pulse  register  circuit,  the  first  three  digits  serving  to  connect  tfee  proper  route  relay  ti©  the 
sender  and  thus  set  the  proper  registering  relays  in  the  sender,  at  which  point  die  decoder 
is  disconnected  from  the  circuit. 

The  sender  is  then  prepared  to  cause  the  district  selector  to  choose  an  interoffice  trunk 
to  an  incoming  frame.  The  district  selectors  are  typical  panel-type  selectors,  except  that 
every  eleventh  terminal  and  an  additional  one  between  the  two  groups  of  five  at  the  top 
of  each  bank  is  arranged  as  an  overflow  terminal  to  stop  the  motion  of  the  selector  when 
all  ten  trunks  below  it  are  busy.  The  trunks  can  be  used  in  larger  groups  than  multiples 
of  10  by  making  the  intermediate  overflow  terminals  busy,  In  which  case  the  selector  will 
pass  over  them,  or  groups  of  five  trunks  are  available  at  the  top  of  each  bank. 

If  450  trunks  are  not  enough  to  provide  the  necessary  trunks  to  all  the  offices  in  the 
exchange  area,  a  group  of  office  selectors  is  employed  to  care  for  all  the  trunks  outgoing 
from  the  office.  The  calls  are  then  routed  by  the  district  selector  to  the  proper  office 
selector  and  thence  to  the  incoming  selector  of  the  called  office. 

The  movements  of  the  selectors  on  both  district  and  office  frames  are  guided  by  a  re- 
verse control  method.  As  the  selector  is  driven  upward  by  the  cork  roll,  it  sends  back 
one  pulse  to  the  sender  each  time  a  brush  moving  on  the  commutator  (at  the  top  of  the 
frame)  makes  contact.  Commutators  for  the  various  types  of  selectors  are  shown  in  Fig. 

21.  The  pulses  are  counted  by  the  sender,  and,  when  the  number  of  pulses  indicates  to 
the  sender  that  the  selector  has  moved  to  the  proper  position,  tiie  sender  opeas  tfee  up- 
drive  magnet  circuit,  stopping  the  selector  motion. 

The  first  selection  on  either  the  district  or  office  frame  involves  the  tripping  of  the  proper 
brush.  If,  as  shown  in  Fig.  20,  the  desired  trunk  appears  on  the  fourth  panel  from  the 


Line  Panel 

Finder  Fr.      Link  Fr. 


Catling 
{Subscriber 


District  Fr. 


Incoming  Fr.        Final  Fr- 


FIG.  20.     Typical  Panel-type  Connection— Panel  Dial  System  (Courtesy  Bell  System) 

bottom,  the  district  selector  will  first  move  up  a  distance  of  four  segments  on  the >,  A  bar  of 
the  commutator  (Fig.  2  in  Fig.  21).  At  that  point  the  selector  will  be  stopped,  the  se- 
quence switch  will  be  turned  two  steps  (from  position  4  to  6  in  Table  1),  changing  tfee 


17-24 


TELEPHONY 


fundamental  circuit  to  prepare  for  the  next  operation,  and  the  trip  rod  will  be  rotated 
into  position  to  trip  the  fourth  brush.  The  selector  is  now  started  upward  again  by  the 
sender,  and  sends  back  a  pulse  for  each  bar  on  the  B  bar  of  the  commutator,  each  of  which 
represents  a  distance  of  ten  trunks.  Thus,  the  upward  movement  of  the  selector  to  the 
proper  trunk  group  is  controlled  by  the  number  of  pulses  sent  to  the  sender  from  the  B 


bar.  When  the  selector  reaches  the  proper  group,  the  sequence  switch  is  again  turned 
two  steps  (position  6  to  8  in  Table  1)  and  the  selector  moves  up  slowly,  testing  each  trunk 
in  the  group  to  find  an  idle  one— a  process  known  as  hunting. 

^•Thoohu^tiug  process  is  common  in  panel  operation.  The  connections  are  shown  in 
7£\T  '-.f  connections  marked  8  being  closed,  since  the  sequence  switch  is  on  position  8 
(I able  1).  If  the  L  relay  is  operated,  the  up-drive  clutch  is  energized  by  current  flowing 
to  ground^nrough  the  L  relay  left-hand  contact,  and  the  selector  moves  up.  As  long  as 


MECHANICAL  SYSTEMS  AND  OPERATION 


17-25 


the  sleeve  contact  of  the  selector  brush  is  grounded,  the  L  relay  is  operated  through  contact 
7  1/2/8-  Since  the  sleeve  contacts  of  busy  trunks  are  grounded  and  those  of  idle  trunks 
are  not  grounded,  the  selector  will  move  a  small  distance  beyond  the  last  busy  trunk. 
However,  even  here  the  L  relay  is  grounded  through  the  commutator  brush  and  the  G  bar 
of  the  commutator.  The  reason  for  this  slight  extra  movement  is  to  allow  the  holding 
pawl  on  the  selector  rack  to  engage  the  correct  slot. 


Sleeve  Terminals 
of  Trunks 


Sleeve  Contact 
of  Tripped  Brush 


FIG.  22.     Selector  Circuit  for  Hunting  Idle  Trunk — Panel  Dial  System  (Courtesy  Bell  System) 

When  an  idle  trunk  is  found,  the  sequence  switch  turns  to  position  10,  until  the  con- 
nection is  completed,  the  changes  necessary  in  the  control  circuits  being  made  by  other 
sequence  switches  located  on  the  incoming  and  final  selector  frames. 

Table  1.    Operations  of  District  Sequence  Switch 


Position 

Corresponding  Circuit  Condition 

Position 

Corresponding  Circuit  Condition 

I 

Normal 

10 

Selection    of    brushes,    groups,    etc.. 

2 

Selecting  an  idle  sender 

beyond  the  district  selector 

3 

Waiting  for  sender 

11 

Waiting  for  sender 

4 

Selecting  brush 

12 

Talking  (non-loaded  trunk) 

5 

Waiting  for  sender 

13 

Talking  (medium-loaded  trunk) 

6 

Selecting  group 

14 

Waiting  for  operator  to  answer 

7 

Waiting  for  relays 

15 

Talking  to  operator 

8 

Hunting  idle  trunk 

16 

All  trunks  busy 

9 

Waiting  for  sender 

17 

Operating  message  register 

18 

Returning  apparatus  to  normal 

These  selectors  are  also  of  the  panel  selector  type,  differing  only  in  details.  The  final 
selector  frame  has  capacity  for  500  lines,  thus  requiring  20  final  frames  for  a  10,000  sub- 
scriber line  office.  The  incoming  selector  frame  must  then  hold  20  groups  of  trunks, 
which  can  consist  of  24  trunks  and  an  overflow  terminal.  This  arrangement,  which  is 
employed  in  all  offices,  is  shown  in  Fig.  23.  The  sender  circuit  is  so  arranged  that  either 
0  or  1  in  the  thousands  digit  causes  brush  0  to  trip,  while  the  combination  of  the  first  two 
numbers  determines  the  proper  group  for  the  selector  to  hunt  in.  When  an  idle  trunk  to 
the  final  selector  is  found,  the  second  or  hundreds  digit  determines  which  brush  will  be 
tripped,  while  the  tens  digit  determines  in  which  group  the  called  line  appears.  Final 
selectors  are  directed  rapidly  to  the  proper  group  of  ten;  when  the  proper  group  is  reached, 
the  clutch  on  the  up-drive  roll  is  released  by  a  change  in  the  setting  of  the  sequence  switch 
and  !/4  usual  speed  is  substituted  as  the  motive  power.  The  final  selector  tests  the  line 
on  which  it  stops,  and  if  it  finds  the  line  busy  it  returns  to  normal  and  sends  back  a  busy 
tone  to  the  calling  subscriber. 

If  the  called  subscriber  has  a  private  branch  exchange  (PBX)  with  several  trunks,  the 
final  selector  will  hunt  for  an  idle  line  if  the  called  number  is  busy  exactly  as  it  hunts  for  an 
idle  interoffice  trunk. 

Many  special  provisions  must  be  made  in  any  installation  for  party  lines,  message  rate 
service,  rapid  testing,  and  other  special  conditions  that  arise,  but  such  arrangements  will 
not  be  considered  here. 


17-26 


TELEPHONY 


THE  CROSSBAR  SYSTEM  is  the  latest  Bell  System  development  in  mechanical  switch- 
ing. It  differs  radically  from  the  Strowger  (step-by-step)  and  the  panel  dial  systems  ^in 
construction,  operation,  and  control,  but  it  is  so  designed  that  it  functions  satisfactorily 
with  all  other  types  of  switching,  whether  mechanical  or  manual.  The  crossbar  system 
offers  important  improvements  in  switching,  both  in  operation  and  maintenance,  over  the 
step-by-step  and  panel  systems,  but  it  does  not  necessarily  replace  existing  installations  or 
additions  to  these  older  systems. 

Only  the  most  important  functions  of  the  crossbar  system  can  be  discussed  in  this 
handbook,  owing  to  space  limitations. 

The  crossbar  system  has  two  outstanding  features,  the  crossbar  switch  which  is  used  for 
all  major  switching  operations,  and  the  marker  system  of  control  which  is  used  in  establish- 
ing all  connections  throughout  the  crossbar  office.  The  apparatus  consists  principally  of 

crossbar  switches  of  the  relay  type  and  multi- 
contact  and  other  type  relays  generally  em- 
ployed in  telephone  systems.  The  switching 
circuits  are  wired  to  the  contacting  springs  of 
the  switches,  and  the  circuit  closures  are  made 
when  the  contacts  are  pressed  together  by  op- 
eration of  the  electromagnets. 

The  use  of  relay-type  apparatus  economically 
permits  having  twin  contacts  of  precious  metal 
throughout,  insuring  reliable  operation  for  the 
low  values  of  speech  and  signaling  currents  in- 
herent to  telephone  systems.  The  very  short 
mechanical  movements  and  small  operating  time 
intervals  required  in  crossbar  switch  operation 
permit  a  reduction  in  control  equipment  over 
the  slower-moving,  older-type  systems,  thus 
resulting,  in  the  use  of  large  switch  and  relay 
assemblies  on  unit-type  frames,  factory  wired 
and  tested.  In  the  design  of  the  switching 
frames  and  associated  control  circuits,  it  has 
been  possible  to  standardize  a  relatively  few 
types  of  equipment  units,  thus  simplifying  manu- 
facture, merchandising,  and  operating  company 
engineering. 

The  marker,  composed  mainly  of  relays  by 
means  of  which  it  controls  switching  operations, 
has  many  advantages,  one  of  the  most  important 


Brush 
4 

3 
2 
1 

0 

Fr 
Ccntr; 

I  neon 

ning 
Group 

Brush 
4 

3 

2 

n 

nal 
Group 

95CO 

9999 

3 

2 
1 
0 

3 
2 
1 
0 

3 
2 

4900 

4999 

4B12g 

9000 

9499 

esoo 

8999 

8000 

8499 

750O 

7999 

4BOO 

4899 

7000 

7499 

650O      |  6999 

600" 

6499 

E50O      !  5053 

47OO 

4799 

fOOO      |  5499 

40CO 

4491 

0 

3 
2         Z 

1      £ 

0            M 

3 
2 
1 

i=n 

»-<  o 
Trunks  to  Final  Selectors 

3SOO 

3000 

4600 

4699 

2CCP 

3499 

250C 

2990 

ccco 

24S9 

1SCO 

1999 

j 

4599 

==^? 

1000 

1499 

050O 

0999 

oooo 

0490 

45fl£~ 

Tsos 

M> 

am 
1  Office 
ing  Lmi 

<&> 

—  II 

1    Trunks  to  Final  Selectors 

on  Final  Frame  for  Lines  0000-0X99 

FIG.  23.    Trunk  and  Subscriber  Multiple  on 

Incoming   and   Final    Selectors — Panel  Dial 

System  (Courtesy  Bell  System) 


being  the  completion  of  its  complex  functional  processes  in  establishing  a  call  in  less  than 
one  second.  The  markers  are  connected  momentarily,  by  means  of  multicontact  relays, 
to  various  switching  units  of  the  office  to  guide  the  completion  of  calls  through  the 
crossbar  switches.  The  marker  system  provides  for  an  attempt  automatically  to  establish 
a  call  over  alternative  paths  when  all  the  normal  routings  are  busy  or  trouble  is  encoun- 
tered; the  marker  will  detect,  record  the  location  and  nature  of  such  troubles,  and  indicate 
their  presence  to  an  attendant  by  operating  an  alarm.  The  marker  design  also  facilitates 
the  introduction  of  new  service  features  and  operating  changes  as  desired,  since  the  main 
control  features  of  the  entire  system  are  incorporated  in  a  small  number  of  markers. 

The  crossbar  switch,  which  gives  the  system  its  name,  is  the  basic  switching  unit  of  the 
system.  Figure  24  shows  a  front  view  of  a  200-point  switch.  Fundamentally,  this 
switch  contains  (1)  10  separate  horizontal  circuit  paths,  (2)  20  separate  vertical  circuit 
paths,  and  (3)  an  electromagnet  for  each  horizontal  and  each  vertical  path,  so  that  the 
operation  of  any  horizontal  and  any  vertical  magnet  in  sequence  will  connect  a  horizontal 
and  a  vertical  path  together  at  one  of  the  200  cross-points.  The  10  horizontal  paths  are 
controlled  by  5  horizontal  bars  each  actuated  by  a  selecting  magnet,  and  the  20  vertical 
paths  are  controlled  by  20  vertical  bars  each  actuated  by  a  holding  magnet.  Any  set  of 
contacts  may  be  closed  by  first  operating  the  selecting  magnet  corresponding  to  the  hori- 
zontal row  in  which  the  contacts  are  located,  and  then  by  operating  the  holding  magnet 
associated  with  the  particular  row  of  vertical  contacts.  The  holding  magnet  will  hold 
the  contacts  closed  until  the  connection  is  released,  but  after  it  is  energized  the  selecting 
magnet  returns  to  normal  until  called  upon  to  operate  on  another  call.  Thus,  10  sets  of 
contacts  may  be  made  at  one  time  through  the  switch,  one  for  each  horizontal  path. 

Figure  25  shows  in  detail  a  portion  of  the  selecting  mechanism  of  the  crossbar  switch. 
The  10  sets  of  contacts  in  each  vertical  row  are  associated  with  the  vertical  or  holding  bar 
of  that  row.  Each  horizontal  or  selecting  bar  has  20  flexible  wire  selecting  fingers,  mounted 


MECHANICAL  SYSTEMS  AND   OPERATION 


17-27 


at  right  angles  to  the  bar,  one  finger  for  each  vertical  row  of  contacts.  When,  a  selecting 
bar  is  rotated  through  a  small  arc  by  its  magnet,  the  selecting  fingers  move  up  or  down  into 
position,  depending  upon  the  direction  of  rotation  of  the  bar.  When  a  holding  bar  in  a 


FIG.  24.     Crossbar  Switch— Front  View — Crossbar  System  (Courtesy  Bell  System) 

particular  vertical  row  is  operated  by  its  magnet,  it  will  bear  against  the  particular  selecting 
finger  which  has  been  moved  into  position  in  its  row,  and  the  finger  will  be  pressed  against 
the  operating  spring  horizontally  in  line  with  it.  Thus,  the  operating  spring  will  be  pressed 
against  the  contact  multiple  (fixed  contact  spring)  and  the  circuit  path  will  be  closed  at 
that  point. 


ADJUSTABLE 
SUPPORT  FOR 
SELECTING 
FINGER 


OPERATING  SPRING 

CONTACT  MULTIPLE 

**A 

ARMATURE 

FIG.  25.     Crossbar  Switch  Selecting  Mechanism — Crossbar  System  (Courtesy  Bell  System) 

The  selecting  bar  and  all  its  fingers  except  the  one  being  held  against  the  operating 
spring  will  be  released  as  soon  as  the  holding  bar  operates.  When  the  connection  is  re- 
leased, the  holding  bar  returns  to  normal  and  the  held  finger  returns  to  its  idle  position 


17-28 


TELEPHONY 


midway  between,  but  slightly  to  the  right  of,  two  sets  of  operating  springs.  At  those 
points  along  an  operated  vertical  bar  where  the  angers  are  not  operated  by  the  horizontal 
bar,  the  fingers  are  pushed  in  between  each  two  sets  of  operating  springs  and  thus  do  not 
bear  against  these  springs.  Only  one  finger  is  operated  at  one  time  for  a  given  vertical 
bar. 

The  selection  operation  is  performed  by  five  horizontal  bars,  each  of  which  will  select 
one  of  two  horizontal  rows  of  operating  springs,  depending  on  the  direction  of  rotation  of 
the  bar  and  consequently  the  selecting  fingers.  Figure  24  shows  two  magnets  at  the  end 
of  each  horizontal  bar,  one  causing  the  fingers  to  move  upward  and  the  other  causing 
them  to  move  downward.  After  release,  the  horizontal  bar  is  restored  to  its  mid  or  idle 
position  by  centering  springs  at  the  ends  of  the  bars  and  adjacent  to  the  magnets. 

The  vertical  units  of  the  crossbar  switch  each  have  10  sets  of  operating  springs  in  vertical 
rows  with  one  vertical  or  holding  magnet  at  the  bottom  of  the  row  which  actuates  one 
vertical  bar  or  armature  extending  the  height  of  the  10  sets  of  contacts.  Each  set  of 
contacts  may  consist  of  three,  four,  five,  or  other  numbers  of  pairs  of  springs  in  horizontal 
spring  pile-up  or  assembly,  depending  on  circuit  requirements.  Figure  24  shows  four 
pairs  of  springs  per  set  of  contacts.  One  twin  contact  spring  of  each  pair  is  stationary  and 
designated  contact  multiple;  the  mate  or  operating  spring  of  the  pair  is  pressed  against 
the  contact  multiple,  when  operated.  This  contact  multiple  spring  is  made  of  one  piece 
of  metal,  insulated  from  the  mounting  and  extending  the  full  length  of  a  vertical  row  of 
contacts.  Wiring  lugs  are  formed  at  the  lower  end  of  these  vertical  metal  pieces,  facing  the 
rear,  to  which  are  wired  the  lines  or  trunks  of  the  vertical  circuit  paths.  On  the  front  and 
at  the  lower  end  of  these  pieces  projections  are  provided  for  testing  purposes.  The  mate 
or  operating  springs  extend  to  the  rear  of  the  switch,  where  wiring  lugs  are  provided  and 
may  be  strapped  horizontally  to  the  corresponding  springs  of  adjoining  vertical  units, 
thus  extending  the  horizontal  paths  across  the  switch  verticals,  or  to  adjoining  switches. 

The  switch  may  have  "off  normal"  contact  spring  assemblies,  if  required,  associated 
with  each  selecting  magnet  which  operates  them  to  perform  various  circuit  functions. 

The  200-point  crossbar  switch  is  9  x/4  i^-  high  and  30V2  in.  long.  A  100-point  switch 
with  10  verticals  is  also  available. 

The  multicontact  relay  used  in  the  crossbar  system  and  shown  in  Fig.  26  resembles  in 
design  the  vertical  unit  of  the  crossbar  switch.  The  relay  is  provided  with  30,  40,  50,  or 


FIG.  26.     Multicontact  Relay — Crossbar  System  (Courtesy  Bell  System) 

60  sets  of  individually  insulated  contacts,  normally  open,  but  closed  when  the  relay  mag- 
nets are  operated.  Each  relay  has  two  separate  magnets,  armatures,  and  associated  groups 
of  springs.  By  operating  the  magnets  independently,  the  unit  can  be  operated  as  two 
separate  relays,  each  closing  15,  20,  25,  or  30  sets  of  contacts,  or  if  both  magnets  are  en- 
ergized in  parallel  the  full  number  of  contacts  of  the  unit  are  closed.  All  contact  springs 
have  twin  contact  surfaces  of  solid  bars  of  precious  metal  because  of  heavy  duty  require- 
ments. This  relay  is  used  mainly  in  the  common  connector  circuits,  where  a  large  num- 
ber of  leads  must  be  connected  simultaneously  to  a  common  circuit. 

"  New  and  improved  general-purpose  small  relays  of  the  XJ  and  Y  types  are  used  in  the 
tsrossbar  system.  These  relays  permit  the  use  of  up  to  24  springs  in  one  assembly,  pro- 


MECHANICAL  SYSTEMS  AND  OPEEATION 


17-29 


viding  for  various  combinations  of  transfer  or  simple  make  and  break  contacts.  The 
springs  are  equipped  with  twin  contacts  of  various  contact  metals,  depending  on  the 
characteristics  of  the  circuits  controlled  by  them.  The  Y  relay  has  a  slow  release  action 
obtained  by  copper  or  aluminum  sleeves  over  the  relay  core. 

CROSSBAR  OPERATION  may  be  more  readily  understood  from  the  block  diagram 
of  the  functional  arrangement  of  the  principal  equipment  units  of  the  system  which  are 
involved  in  a  connection  between  two  subscribers,  as  shown  in  Fig.  27.  The  three  main 


FIG.  27.     Functional  Arrangement  of  Equipment  Units — Crossbar  System  (Courtesy  Bell  System) 

types  of  equipment  units  are:  (1)  the  district  junctors  and  incoming  trunks,  which  supply 
battery  to  the  transmission  and  supervisory  circuits;  (2)  the  crossbar  switch  frames;  (3) 
the  common  control  circuits  and  the  senders  and  markers. 

The  district  junctor  and  incoming  trunk  circuits  are  composed  mainly  of  small  relays. 
The  district  junctor  furnishes  talking  battery  for  the  calling  subscriber  and  supervises 
the  originating  end  of  the  connection.  The  incoming  trunk  circuit  controls  the  ringing  of 
the  called  subscriber  bell,  furnishes  talking  battery  for  the  called  line,  and  supervises  the 
terminating  end  of  the  connection. 

The  switch  frames,  consisting  principally  of  crossbar  switches,  provide  the  means  for 
switching  between  the  subscriber  lines,  district  junctors,  and  incoming  trunks,  and  also 
for  switching  these  district  junctors  and  trunks  to  senders. 

The  senders  consist  chiefly  of  small  relays,  and  their  function  in  crossbar  is  comparable 
to  that  of  an  operator  in  manual  operation.  The  subscriber  sender  registers  the  called 
number  from  the  subscriber  dial  pulses  and  transmits  the  necessary  information  to  the 
markers,  to  the  terminating  sender,  and  to  the  manual  operators  (in  manual  offices)  for 
completing  connections  to  the  called  line.  This  sender  also  controls  operation  of  the 
selectors  in  distant  panel  offices.  The  terminating  sender  receives  the  numerical  digits 
of  the  called  number  from  the  subscriber  sender  of  any  panel  or  crossbar  office  and  trans- 
mits the  required  information  to  the  terminating  marker  for  setting  up  connections  to  the 
called  line. 

The  markers  are  the  most  important  control  circuits  in  the  crossbar  system.  They 
comprise  both  small  and  multieontact  relays  and  are  of  two  types,  one  for  originating  and 
one  for  terminating  traffic.  Since  their  operating  time  is  less  than  1  sec,  only  three  or 
four  markers  are  needed  in  the  average  size  office. 

The  originating  marker  determines  the  proper  trunk  routes  to  the  called  office.  It  has 
access  to  all  outgoing  trunk  circuits  and  all  crossbar  switch  frames  used  in  establishing 
connections  to  the  called  office  trunks.  The  marker  records  pulse  information,  tests  the 


17-30 


TELEPHONY 


trunk  group  for  an  idle  trunk  to  the  called  office,  tests  for  and  marks  or  reserves  an  idle 
path  through  the  switch  frames,  and  finally  operates  the  proper  crossbar  switches  to  es- 
tablish a  path  from  the  calling  subscriber  line  to  the  outgoing  trunk  circuit.  The  lowest- 
numbered  available  paths  are  always  selected  in  order  to  reduce  selection  time  and  in- 
crease operating  efficiency. 

Trunk  selection  is  made  by  the  marker  through  route  relays,  of  which  there  is  one  for 
each  called  office  routing.  This  relay  is  so  wired  as  to  direct  the  marker  to  the  called 
office  trunk  group  and  indicate  the  number  of  trunks  in  the  group;  also  to  indicate  the 
office  link  switch  frame  on  which  the  trunk  group  appears  and  the  type  of  called  office, 
which  is  also  indicated  to  the  sender.  Route  relays  may  be  assigned  to  any  office  trunk 
group,  and  other  changes  may  be  made,  as  required. 

The  terminating  marker  performs  similar  functions  in  the  terminating  office,  establish- 
ing a  path  between  the  incoming  trunk  circuit  and  the  called  subscriber  line.  This  marker 
has  access  to  all  the  subscriber  lines  terminating  in  the  office  and  to  all  crossbar  switch 
frames  used  for  connecting  to  subscriber  lines.  It  records  pulse  information,  tests  the 
called  line  to  determine  whether  it  is  idle,  tests  for  and  marks  an  idle  path  through  the 
switch  frames,  and  finally  operates  the  proper  crossbar  switch  magnets  to  establish  con- 
nection to  the  called  line. 

The  marker,  in  testing  and  connecting  the  called  line,  employs  a  marker  group  con- 
nector circuit  and  block  relay  frame  in  which  the  called  line  appears.  Each  subscriber 
line  has  three  test  terminals  on  the  block  relay  frame,  and  a  number  group  connector  will 
usually  have  access  to  the  test  terminals  of  several  hundred  lines.  The  marker  deter- 
mines from  these  test  terminals  whether  the  called  line  is  busy  and  the  identification  of  the 
proper  line  link  frame  and  horizontal  group  of  line  links  which  have  access  to  the  called 
line;  also  the  type  of  ringing  required  is  determined  from  circuit  conditions  on  the  test 
terminal. 

There  are  also  common  control  circuits,  associated  with  the  line  link  and  sender  link 
frames,  for  controlling  the  operation  of  the  switches  on  these  frames.  In  addition,  there 
are  common  connector  circuits,  composed  mainly  of  multicontact  relays,  which  are  used  to 
connect  the  markers  (1)  to  their  respective  senders,  (2)  to  their  associated  switch  frames, 
and  (3)  to  the  subscriber  line  test  terminals. 

The  line  link  frames,  although  shown  separately  in  Fig.  27,  are,  in  a  given  office,  used 
for  both  originating  and  terminating  traffic.  After  the  talking  connection  has  been  es- 
tablished between  subscribers  (see  Fig.  28) ,  all  the  common  control  circuits,  including  send- 
ers, markers,  connectors,  line  link  control  circuits,  sender  link  frames,  and  their  associated 


CALLING 
TELE- 
PHONE 

a 

LINE 
LINK 
FRAME 

i~r 

DISTRICT 
JUNCTOR 
CALLING 
TELEPHONE 
TRANS- 
MISSION AND 
SUPERVISORY 
RELAY 
EQUIPMENT 

DISTRICT 
LINK 
FRAME 

«rt. 

OFFICE 
LINK 
FRAME 

•%J- 

INCOMING 
TRUNK 
CALLED 
TELEPHONE 
TRANS  - 
"      MISSION,     " 
SUPERVISORY 
RELAY,  AND 
RINGING 
EQUIPMENT 

INCOMING 
LINK 
FRAME 

LINE 
LINK 
FRAME 

4o-of 

CALLED 
TELE- 
PHONE 

o 

FIG.  28.     Completed  Talking  Connection — Crossbar  System  (Courtesy  Bell  System) 

control  circuits,  will  have  been  released,  and  the  talking  path  will  be  maintained  by  the 
holding  magnets  of  the  crossbar  switches,  which  are  used  on  the  link,  district,  office,  and 
incoming  link  switch  frames.  These  holding  magnets  are  held  operated  under  control  of 
the  supervisory  relays  in  the  district  junctor  and  the  incoming  trunk  circuits  and  are  re- 
leased only  when  the  subscribers  hang  up  their  handsets. 

The  establishment  of  a  connection  with  the  crossbar  switch  is  shown  schematically  in 
Fig.  29T  in  which  20  vertical  units  are  connected  to  20  subscriber  lines  and  10  trunks  are 
strapped  horizontally  across  the  switch.  With  this  arrangement,  any  one  of  the  20  lines 
may  be  connected  to  any  one  of  the  10  trunks  by  closing  the  contacts  at  the  proper  cross 
point.  By  adding  a  second  200-point  switch  with  20  additional  lines  connected  to  its 
verticals,  and  extending  the  trunk  strapping  through  both  switches,  40  lines  are  given 
access  to  the  10  trunks.  Thus,  by  adding  other  switches  in  this  manner,  the  number  of 
lines  having  access  to  these  10  trunks  may  be  further  increased. 

A  line  link  frame  comprises  primary  bays  and  secondary  bays.  Each  primary  bay 
terminates  200  subscriber  lines  (10  primary  switches  with  20  lines  each),  but  the  number 
of  primary  bays  per  frame  may  be  varied  within  limits  to  meet  traffic  requirements.  The 
secondary  bay  contains  secondary  switches;  the  bay  is  divided  vertically  in  the  center, 
so  that  there  are  10  switches,  each  with  10  verticals  on  the  left  of  center  and  the  same 
arrangement  on  the  right  of  center  of  the  bay.  The  switches  on  the  left  have  their  ver- 


MECHANICAL  SYSTEMS  AND  OPERATION 


17-31 


TO  ro 

TRUNKS 


ticals  connected  to  line  junctors  which  are  used  for  terminating  traffic,  and  those  on  the 

right  have  their  verticals  connected  to  district  junctors  which  are  used  for  originating 

traffic.     At  the  bottom  of  the  secondary  bay  is  a  cabinet  containing  control  circuit  relays, 

and  just  above  this  cabinet 

are  the  multicontact  relays        _  FROM  20  LINES 

which  connect  the  control 

circuits     to     the     crossbar 

switches. 

Since  each  subscriber  in 
an  office  has  only  one  cross- 
bar appearance  and  that  on 
a  vertical  unit  of  a  primary 
crossbar  switch,  both  origi- 
nating and  terminating  calls 
are  completed  by  means  of 
the  same  line  link  circuits 
serving  that  particular 
switch.  Thus,  all  originat- 
ing traffic  from  any  of  the 
20  lines  on  a  primary  switch 
flows  through  the  associated 
10  line  links  to  the  100  district  junctors,  and  all  terminating  traffic  to  these  20  lines  flows 
through  the  same  10  line  links  from  the  line  junctors,  as  shown  in  Fig.  30. 

The  arrangement  shown  in  Fig.  30  is  also  used  in  the  originating  and  terminating  sender 
link  switch  frames,  where  the  circuits  reached  are  non-directional,  that  is,  where  any  one 
of  the  circuits  wired  to  the  frame  and  available  for  selection  can  be  used  for  establishing  a 
connection. 

Where  it  is  necessary  to  provide  greater  flexibility  and  efficiency  in  trunk  groups  than 
is  possible  with  the  arrangement  shown  in  Fig.  30,  two  primary  and  secondary  switch 


)   i   ; 

>  - 

i  i 

\   i 

>  « 

T  e 

5    ? 

1 

D   I 

! 

. 

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i  u 

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>  i 

S   17  18  19 

tit  .., 

1 

1          >  n 

1     - 

\ 

" 

fc     -5 

»     -1 

! 

I 

1 

°j 

FIG.  29.     Simple  Trunking  Arrangement  with  a  Single  200-point 
Crossbar  Switch — Crossbar  System  (Courtesy  Bell  System) 


SUBSCRIBER 
LINES 


DISTRIBUTING 
FRAME 


FRAME    SERVES 
ISO  TO  70O  LINES 
DEPENDING  ON 
TRAFFIC 


TO 

DISTRICT 
JUNCTOR 
FRAME 


FRAME  HAS 
ACCESS  TO 
100  DISTRICT  JUNC- 
TORS, 100  LINE 
JUNCTORS 


PRIMARY  SECONDARY 

FIG.  30.    Single  Primary-secondary  Trunking  Arrangement — Crossbar  System  (Courtesy  Bell  System) 


frames  are  employed,  as  shown  in  Fig.  31.  This  layout  shows  an  incoming  link  frame  to 
which  incoming  trunks  are  connected,  and  a  line  link  frame  to  which  subscriber  lines  are 
connected.  These  two  frames  are  operated  in  tandem  for  establishing  the  terminating 
connections  between  the  incoming  trunks  and  are  called  subscriber  lines.  As  shown,  100 
incoming  trunks  are  connected  to  the  100  horizontal  paths  of  the  10  incoming  link  frame 
primary  switches,  10  trunks  per  switch.  Although  only  200  lines  (20  lines  on  each  of  10 


17-32 


TELEPHONY 


primary  switches)  are  indicated  in  Fig.  31,  150  to  700  subscriber  lines  mav  appear  on  the 
verticals  of  the  primary  switches  of  the  line  link  frame. 

Referring  to  Fig.  31,  the  connection  of  a  particular  incoming  trunk  to  a  particular  called 
line  requires  the  selection  of  an  idle  path  through  the  incoming  link  and  line  link  frames. 
This  path  will  consist  of  an  incoming  link,  a  line  junctor,  and  a  line  link.  The  incoming 
trunks  on  each  of  the  primary  switches  have  access  to  20  incoming  links  appearing  on  the 
20  verticals  of  the  switch.  These  20  incoming  links  are  distributed  over  the  10  secondary 
switches  of  the  incoming  link  frame,  two  links  to  a  switch  and  one  link  to  each  half  switch. 
In  order  to  provide  for  the  distribution  of  the  20  incoming  links  over  the  10  secondary 
switches,  the  horizontal  paths  of  the  secondary  switches  are  separated  between  the  tenth 
and  eleventh  verticals,  thus  providing  20  instead  of  10  horizontal  paths  on  each  switch. 
The  incoming  links  on  each  half  of  these  secondary  switches  have  access  to  line  junctors 
appearing  on  the  verticals  of  these  switches.  These  junctors  are,  in  turn,  distributed  over 


INCOMING  LINK  FRAME 
PRIMARY  SWITCH  9  SECONDARY  SWITCH  9 


LINE  LINK  FRAME 
SECONDARY  SWITCH  9        PRIMARY  SWITCH  9 


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10  DISTRICT 
JUNCTORS 

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vj_       '  , 
TO  RIGHT  HALF 
OF   SECONDARY 
SWITCHES 

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JUNCTORS              INCOMING  — 

LINK  FRAME 
FIG.  31.    Double  Primary-secondary  Trunking  Arrangement — Crossbar  System  (Courtesy  Bell  System) 

the  secondary  switches  of  all  the  line  link  frames  in  the  office.  There  will  be  at  least  one 
junctor  from  each  secondary  switch  on  an  incoming  link  frame  to  a  secondary  switch  on 
every  line  link  frame  in  the  office,  or  a  minimum  of  10  line  junctor  paths  between  any  in- 
coming link  frame  and  any  line  link  frame,  the  number  of  paths  varying,  depending  on  the 
number  of  frames  required  in  an  office. 

The  line  junctors  on  the  verticals  of  each  of  the  line  link  frame  secondary  switches  have 
access  to  10  line  links  on  the  horizontal  paths.  The  10  line  links  are  distributed  over  the 
primary  switches  of  the  line  link  frame,  one  to  each  switch,  giving  each  link  access  to  the 
called  subscriber  lines  appearing  on  the  verticals  of  the  primary  switch  with  which  the 
link  is  associated. 

With  the  Fig.  31  arrangement  of  switches  and  the  three  groups  of  interconnecting  link 
paths,  any  incoming  trunk  can  be  connected  to  any  called  line  appearing  on  the  line  link 
frame  or,  by  means  of  other  groups  of  line  junctors,  to  a  called  line  on  any  other  line  link 
frame  in  the  office.  This  trunking  arrangement  is  also  employed  for  connecting  district 
junctors  to  outgoing  trunks  in  the  originating  office. 

Establishing  a  call  from  one  crossbar  subscriber  to  another  crossbar  subscriber  requires 
four  stages  of  operation,  two  at  the  originating  and  two  at  the  terminating  office: 

1.  The  calling  subscriber  is  connected  to  a  subscriber  sender  through  the  line  link  frame, 
district  junctor,  and  sender  link  frame,  and  the  sender  registers  the  dial  pulses  of  the  called 
number. 

2.  The  subscriber  sender  is  connected  to  an  originating  marker  through  the  marker 
connector,  and  the  marker  then  selects  the  switch  frames  for  establishing  a  connection  be- 
tween the  calling  subscriber  line  and  an  outgoing  trunk. 

3.  The  outgoing  trunk  (uicoming  at  the  terminating  office)  is  connected  to  a  terminating 
sender  through  the  terminating  sender  link  frame,  and  the  sender  registers  the  pulses  cor- 
responding to  the  called  number. 


MECHANICAL  SYSTEMS  AND   OPERATION 


17-33 


4.  The  terminating  sender  is  connected  to  a  terminating  marker  through  a  marker  con- 
nector, and  the  marker  then  selects  the  switch  frames  for  establishing  the  connection  to 
the  called  subscriber  line. 

Future  developments  in  crossbar  equipment  point  toward  simplification  of  equipment  and 
circuits  and  reductions  in  the  number  of  units  required,  wherever  possible.  An  improved 
system  of  crossbar  is  now  under  study,  in  which  it  may  be  possible  to  establish  connections 
with  one  marker  instead  of  the  two  markers  now  used,  and  in  which  other  simplifications 
may  be  secured. 

OTHER  RELAY  SWITCHING  SYSTEMS  developed  by  various  manufacturing 
companies  include  Relaymatic,  by  Kellogg  Switchboard  and  Supply  Co.;  Relaydial,  by 
Stromberg-Carlson  Co.;  All-Relay,  by  North  Electric  Manufacturing  Co. 

These  systems  operate  on  the  principle  of  (1)  finding  the  calling  line  when  the  calling 
subscriber  takes  his  handset  from  the  hook,  (2)  selecting  and  closing  groups  of  contacts 
by  relay  action  under  control  of  dial  pulses,  subdividing  the  groups  of  contacts  closed, 
until  the  contacts  of  the  called  line  are  reached,  after  which  ringing  power  is  applied  to  the 
called  line  by  a  link  circuit.  Since  no  moving  parts  are  involved  except  the  operating 
springs  or  reeds,  which  are  equipped  with  precious-metal  twin  contacts,  maintenance  is 
simplified  and  operating  costs  are  reduced  over  other  types  of  automatic  switching  employ- 
ing motors  and  step-by-step  type  switches  with  base-metal  contacts  and  sliding  brushes. 

The  line  circuits  may  be  assigned  for  common-battery  local  and  rural,  trunk  or  pay 
station  service.  Local  lines  may  have  individual  stations  or  multiparty  service  up  to 
10-party  selective  (for  metallic  lines)  or  16-  to  20-party  code  ringing  (for  grounded  lines). 
Local  lines  are  of  the  metallic  type,  and  line  adapters  are  used  for  grounded  rural  lines,  as 
required.  By  the  addition  of  a  trunk  adapter,  any  line  circuit  may  be  converted  to  a  trunk 
circuit.  All  link  (connector)  circuits  have  access  to  all  lines,  are  assigned  in  rotation,  and 
automatically  release  from  any  line 
in  trouble  or  as  soon  as  the  subscrib- 
ers hang  up.  A  line  lockout  feature 
is  generally  provided  with  these  sys- 
tems which  prevents  tie-up  of  a  link 
equipment  if  a  line  is  in  trouble  or  a 
handset  is  left  off  the  hook  or  if  the 
link  fails  to  release  after  a  predeter- 
mined time.  As  soon  as  the  line  in 
trouble  is  restored  to  normal,  the 
lockout  of  the  line  from  access  to  the 
link  circuits  is  automatically  discon- 
tinued. 

These  systems  are  made  in  capaci- 
ties from  10  to  10,000  lines  and  may 
be  operated  as  unattended  small  dial 
offices,  if  desired,  with  trunks  to  a 
nearby  manual  or  mechanical  office. 
In  such  cases,  suitable  alarms  are 
provided  which  indicate  at  the  at- 
tended office  when  the  equipment 
needs  attention  and  the  type  of 
trouble  at  the  unattended  office. 
Some  systems  of  more  than  200  lines 
employ  relay-type  selectors  to  dis- 
tribute calls  from  one  group  of  100 
line  finders  to  the  desired  100-Hne 
group  of  connector  links  and  con- 
nectors. 

McBerty  Automatic  Telephone 
System  (North  Electric  Manufactur- 
ing Co.).  The  important  unit  of 
this  system  is  the  McBerty  relay, 
Fig.  32,  a  new  design  consisting  of 
an  integral  reed  spring-armature-contact  structure  having  no  pivots  or  hinges,  which  are 
common  to  most  relays.  The  basic  mounting  structure  may  be  used  as  a  single  multi- 
contact  relay  or  as  a  group  of  three,  four,  or  five  separate  relays,  the  entire  unit  being 
light  and  compact.  The  relay  coils  are  of  the  molded  bobbin,  type,  which  slip  over  the 
steel-alloy  cores  welded  to  the  mounting  frame.  Gold-alloy  contacts  are  used  throughout. 
Bare  wire  is  used  for  multipling  relay  contacts. 


FIG.  32.    McBerty  Relay  (Courtesy  North  Electric  Mfg. 
Co.) 


17-34 


TELEPHONY 


This  system  employs  a  link  circuit  consisting  of  a  line  finder,  connector  control  relays, 
and  a  connector  for  establishing  a  connection  between  a  calling  and  called  subscriber.  As 
many  links  are  provided  as  are  required  to  handle  normal  traffic  loads.  Figure  33  shows 
a  single  link  circuit  layout  for  a  100-line  system,  but  for  purposes  of  illustration  only 
three  of  the  ten  tens  relays  are  shown.  Also  for  clarity  each  single  line  shown  represents 
two  wires  outside  of  and  three  wires  within  the  switchboard. 

When  a  calling  subscriber  removes  his  handset  from  the  hook,  the  line  relay  in  his  line 
circuit  operates,  causing  the  proper  line  finder  tens  and  units  relays  of  an  idle  link  circuit 
to  operate,  closing  the  calling  line  through  to  the  connector  control  relays  of  this  link. 
The  calling  subscriber  then  dials  the  called  number,  and  the  dial  pulses  are  registered  by 


LINES 


LINE -FINDER  <1,WF1  CW<!)  CONNECTOR 

UNITS    RELAYS 

PIG.  33.     Diagram  of  North  100-line  All-relay  System  (Courtesy  North  Electric  Mfg.  Co.) 

the  control  relays.     These  relays  cause  operation  of  the  connector  tens  and  units  relays 
(corresponding  to  the  called  number)  connecting  the  called  line  to  this  link. 

The  above  operations  complete  the  connection  between  the  calling  and  called  subscribers 
through  the  link  circuit,  which  applies  ringing  power  to  the  called  line.  This  link  is  not 
released  for  other  connections  until  the  subscribers  hang  up.  While  the  tens  relays  close 
through  ten  lines  in  both  the  line  finder  and  connector  each  time  they  are  operated,  only 
one  units  relay  is  closed  through  in  each  line  finder  and  connector  for  a  given  caU,  and  thus 
all  lines  except  the  calling  and  called  lines  remain  open  to  the  control  relays  and  may  be 
seized  by  other  links  when  calls  are  originated  by  such  lines. 

The  XY  dial  telephone  system,  developed  by  Stromberg-Carlson  Co.,  employs  for  its 
basic  unit  the  XY  selector-type  switch,  a  view  of  which  is  shown  in  Fig.  34. 

The  XY  switch,  used  in  the  line  finder,  selector,  and  connector  circuits  is  radically 
different  in  construction  from  the  Strowger  step-by-step  switch.  The  switch  is  assembled 
on  a  metal  base  plate,  which  is  mounted  horizontally  on  a  switch  frame.  The  switch  has 
a  carnage  with  four  separate  wipers,  tip,  ring,  sleeve,  and  hunt,  and  the  carriage  is  moved 
as  a  unit  in  two  horizontal  directions,  one  paralleling  the  front  edge  of  the  base  plate  called 
the  X  direction,  and  the  other  at  right  angles  to  the  X  direction,  called  the  Y  direction 
under  control  of  two  magnets,  X  and  Y  respectively.  The  carriage  is  driven  by  a  cog  roUer 
(tubular  shaft)  assembly,  which  slides  along  a  shaft  during  the  X  motion  and  rotates  dur- 
ing the  Y  motion.  The  cog  roller  is  a  double-cut  tubular  gear,  with  ratchet  teeth  cut  par- 
allel to  its  length  and  rack  teeth  cut  as  rings.  The  annular  rack  teeth  mesh  with  and  are 


MECHANICAL  SYSTEMS  AND  OPERATION 


17-35 


driven  in  the  X  direction  by  a  sprocket  actuated  by  the  X  magnet.  The  Y  magnet  ac- 
tuates a  pawl  which  engages  the  ratchet  teeth  of  the  cog  roller,  turning  it  around  the 
shaft. 

Since  the  switch  is  100  point,  the  X  motion  is  given  10  steps  (plus  one  for  overtravel) 
and  the  Y  motion  10  steps  (plus  one  for  overtravel),  thus  providing  for  selection  of  any 
one  of  100  lines.  Since  four  wipers  are  involved  and  each  wiper  has  its  own  set  of  11  wire 
banks  (each  11  wires  deep),  44  rows  of  wires  are  lined  up  in  front  of  the  wipers.  An  X 
wiper  is  also  provided,  which  is  operated  by  a  pinion  and  rack  assembly  actuated  by  the 
X  magnet  and  which  has  access  to  a  23-wire  bank  to  mark  the  level  of  X  travel. 

When  the  X  wiper  finds  the  proper  level,  thus  positioning  the  wipers,  on  the  carriage, 
before  the  proper  wire  banks,  the  X  magnet  is  de-energized  and  the  Y  magnet,  assuming 
control,  moves  the  wipers  into  the  wire  banks  until  the  proper  line  wires  are  reached. 


Carriage 


Sleeve 
and- 
Hunt 


-^-Mechanism 
Plate 


X  Magnet 


•Y  Magnet 


•Release 
Magnet 


FIG.  34.     The  XY  Switch  (Courtesy  Stromberg-Carlson  Co.) 

A  number  of  unique  features  are  built  into  the  XY  switch  assembly,  such  as  bare  wire 
multiple  for  line  terminations,  a  new  mechanical  design  for  magnet  current  interruptions 
to  avoid  armature  chatter,  and  flexibility  to  function  as  a  line  finder,  selector,  or  con- 
nector; the  fact  that  the  common  wire  banks  need  be  wired  only  once,  for  up  to  50  switches, 
results  in  large  wiring  economies. 

The  system  requires  line  finders,  selectors,  and  connectors,  as  in  other  step-by-step 
systems,  but  the  XY  switch  functions  for  each  of  these  three  units. 

In  operation,  the  calling  subscriber  removes  his  handset  from  the  hook,  thereby  causing 
his  line  relay  to  operate  and  an  idle  line  finder,  which  is  permanently  associated  with  a 
selector,  to  find  and  connect  to  the  calling  line.  The  calling  line  is  thus  extended  through 
to  a  selector  which  returns  dial  tone  and  supplies  talking  battery  to  the  calling  subscriber. 
Assuming  that  the  called  number  is  234,  the  first  digit  dialed  is  2,  which  causes  the  selector 
to  move  two  steps  in  the  X  direction  and  into  its  wire  bank  automatically  in  the  Y  direc- 
tion until  an  idle  connector  serving  the  200  group  of  lines  is  found.  When  the  second 
digit,  3,  is  received  by  the  connector,  it  moves  its  wipers  three  steps  in  the  X  direction,  and 
when  the  third  digit,  4,  is  received,  these  wipers  move  into  the  wire  banks  in  the  Y  direc- 
tion to  the  fourth  wire,  thus  connecting  the  calling  to  the  called  line.  This  connection  is 
not  completed,  however,  until  the  connector  applies  ringing  power  to  the  called  line  and 
the  called  subscriber  has  answered. 


17-36  TELEPHONY 


3.  TOLL  SYSTEMS  AND  OPERATION 

Toll  systems,  as  distinguished  from  local  systems,  are  designed  to  handle  toll  traffic 
over  toll  circuits.  Toll  traffic  differs  very  materially  from  local  traffic,  since  toll  circuits 
may  extend  from  one  toll  office  to  a  nearby  toll  office  or  to  an  office  in  this  country  or  in 
almost  any  other  country  in  the  world,  involving  many  thousands  of  miles  of  circuit. 

ToH  circuits  must  be  of  a  grade  suitable  in  all  respects  for  the  traffic  they  are  required 
to  handle.  For  the  very  long  circuits,  expensive  equipment  and  complex  arrangements 
are  necessary  to  meet  all  requirements. 

In  the  smaller  offices  the  toll  and  local  positions  are  identical  or  in  the  same  lineup;  the 
same  line  and  cord  circuits  may  be  used  for  both  local  and  toll  service.  In  the  larger  toll 
centers,  involving  a  number  of  toll  circuit  groups,  separate  manual  toll  boards  are  provided 
for  concentrating  in  one  switchboard,  for  a  given  toll  area  or  center,  all  the  toll  circuits 
serving  that  area.  Such  switchboards  require  special  circuits  and  auxiliary  equipment  for 
properly  recording,  ticketing,  timing,  and  supervising  toll  connections. 

MANUAL  TOLL  SWITCHBOARDS  now  in  use  in  large  metropolitan  centers  provide 
outward  positions  for  outgoing  toll  calls,  inward  positions  for  incoming  toll  calls,  and 
through  positions  for  toll  calls  switched  through  the  board  from  one  toll  circuit  to  another, 
or,  if  desired,  toll  positions  may  be  designed  to  handle  both  inward  and  through  toll  calls. 

One  type  of  manual  toll  switchboard  (Western  Electric  Co.  No.  3C)  now  in  use  is 
equipped  to  handle  all  types  of  manual  toll  switching  and,  in  connection  with  step-by-step 
offices,  may  be  arranged  to  handle  those  local  dial  calls  requiring  the  assistance  of  an 
operator. 

The  functions  of  this  particular  type  of  board  are  (1)  to  establish  outward  toll  con- 
nections while  holding  the  calling  subscriber  on  the  line  (combined  line  and  recording 
[CLR]  traffic) ,  or  to  establish  connections  later,  if  for  any  reason  the  call  cannot  be  com- 
pleted on  the  first  attempt;  (2)  to  connect  inward  toll  calls  to  the  called  subscriber  line 
if  in  the  local  or  tributary  office  area;  (3)  to  interconnect  toll  circuits  (through  traffic) 
upon  request  of  a  distant  operator;  and  (4)  to  handle  miscellaneous  local  calls. 

Outgoing  calls  from  local  or  tributary  (small  office,  having  toll  connections  to  its  larger 
toll  center  office)  offices  reach  the  toll  board  over  recording-completing  trunks  from  local 
manual  or  mechanical  offices  or  over  tributary  toll  circuits  from  tributary  offices.  Inward 
calls  reach  the  toll  operator  over  toll  circuits  (between  toll  centers)  and  are  completed  to 
the  local  called  subscribers  over  toll  switching  trunks,  either  on.  a  straightforward  basis 
through  manual  B  boards  or  by  dialing  or  key  pulsing  over  mechanical  trunks  through  a 
mechanical  office. 

This  switchboard  has  nine  jack  panels  and  three  operator  positions,  each  equipped  with 
ten  pairs  of  high-impedance  toll  cord  circuits,  per  section,  and  by  adding  sufficient  operator 
positions  to  handle  the  traffic  load  as  many  toll  and  trunk  circuits  as  required  can  be  ter- 
minated in  the  switchboard. 

These  boards  are  equipped  with  calculagraphs  for  stamping  the  tickets  with  the  elapsed 
time  of  calls  for  billing  purposes,  electric  clocks,  ticket  conveyors,  ticket  holders  and  com- 
partments, and  many  other  auxiliary  devices  for  handling  toll  traffic.  Transmission  gain 
may  be  introduced  in  the  toll  circuits,  as  required,  when  the  operator  inserts  the  plug  of 
her  cord  circuit  into  the  toll  line  jack.  Idle  circuits  and  circuits  busy  may  both  be  in- 
dicated by  lamp  signals,  and  automatic  listening  equipment  may  be  provided  at  inward 
positions  for  incoming  plug-ended  toll  circuit  operation. 

MECHANICAL  TOLL  SWITCHBOARDS  OR  SYSTEMS  have  been  in  use  for  some 
years  in  a  number  of  comparatively  small  toll  networks,  the  first  systems  being  of  the  dial 
or  step-by-step  type, 

The  step-by-step  system  of  toll  dialing,  though  useful  and  economical  for  a  small  group 
of  interconnected  dial  offices,  presents  sizable  operating  problems  where  intertoll  dialing 
is  attempted  over  an  extensive  area  involving  a  large  number  of  intermediate  offices. 

The  mechanical  switching,  of  toll  traffic  requires  that  the  digits  dialed  by  a  calling  sub- 
scriber or  originating  operator  at  a  given  office,  to  reach  a  particular  subscriber  in  another 
part  of  the  country,  must  be  different  from  the  digits  dialed  to  reach  any  other  subscriber. 
In  step-by-step  toll  dialing  offices,  the  originating  toll  office  is  reached  by  the  subscriber 
dialing  zero  (0),  tributary  offices  by  dialing  the  figure  one  (1),  and  two  more  digits  are 
generally  required  to  select  the  proper  outgoing  toll  or  tributary  circuit.  Thus,  when  a 
call  is  dialed  through  a  number  of  step-by-step  toll  offices,  three  digits  must  be  dialed  for 
each  office  passed  through  in  addition  to  the  digits  required  for  the  terminating  office  and 
called  subscriber  line.  This  requirement  results  in  a  long  series  of  dialed  digits  for  a  call 
that  passes  through  a  number  of  intermediate  offices.  Figure  35  shows  that  19  digits  are 
required  for  a  call  from  Portsmouth,  N.  H.,  to  Vinland,  Kans.,  involving  only  four  inter- 


TOLL  SYSTEMS  AND  OPERATION 


17-37 


mediate  offices.  Additional  intermediate  offices,  in  this  connection,  would  increase  the 
digits  to  be  dialed,  so  that  not  only  would  there  be  dela^"  in  ascertaining  the  proper  codes 
to  dial  in  order  to  extend  the  call  through  the  various  intermediate  offices,  but  also  the 
dialing  of  a  long  series  of  numbers  would  retard  operating  time,  would  hold  expensive 
facilities  unnecessarily  long,  and  would  tend  to  increase  operating  errors.  Also,  if  delays 


PORTSMOUTH,        BOSTON 
N.H.  053 


NEW  YORK 
062 


ST.  LOUtS 
078 


KANSAS   CITY 
O26 


LAWRENCE 

133 


jf fl T 


VINLAND,  KAN. 
1234 


FIG.  35.     Intertoll  Dialing  Scheme — Step  by  Step  System  (Courtesy  Bell  System) 

were  encountered  in  securing  any  intermediate  link  or  if  the  called  line  ras  busy,  the 
complete  dialing  process  would  have  to  be  attempted  again. 

THE  NO.  4  CROSSBAR  TOLL  SYSTEM,  a  development  of  the  Bell  System,  was  first 
placed  in  service  in  Philadelphia  in  August  1943.  This  system  was  developed  primarily 
to  provide  for  ultimate  toll  dialing  on  a  nationwide  basis  and  was  designed  so  that  it  could 
be  introduced  gradually  throughout  the  country  on  an  economical  basis  without  im- 
mediately displacing  existing  manual  or  mechanical  systems  except  as  desired. 

The  No.  4  system  is  arranged,  as  shown  in  Fig.  36,  to  complete  (1)  outward  calls  from 
local  subscribers  to  outgoing  toll  lines,  (2)  inward  calls  from  incoming  dial  or  manual  lines 


\ 

TO  DISTANT  TOLL  OFF 

t 

SUBSCRIBERS 

TRAFFIC   FROM   SUBSCRIBERS  IN  PHILADELPHIA  TOLL  CENTER  AREA  TO  OTHER  TOLL  AREAS 


LOCAL 

OFFICE 


DSA  OR  OUTWARD 
TOLL  BOARD 


CROSSBAR 
EQUIPMENT 


INCOMING  DIAL  TOLL  LINE 


-*• 


INCOMING  RINGDOWN    OR 
STRAIGHTFORWARD  TOLL  LINE 


L±D 


LOCAL 

OFFICE 


SUBSCRIBERS 


LOCAL 
OFFICE 


SUBSCRIBERS 


OPERATOR  POSITION 
(CONNECTED  TO  CONTROL 

CROSSBAR  EQU»PM€NT} 
TRAFFIC  TERMINATING  IN    PHILADELPHIA  TOLL  CENTER  AREA 


INCOMING  DIAL  TOLL  LINE 


OUTGOING  DIAL  OR 


RJNGDOWN  TOLL  LINE 


STRAIGHTFORWARD  TOLL 

LINE 

_                 OUTGOING  DIAL  OR 

1 

RI  NGDOWN  TOLL  LI  N  E 

C±3 


CROSSBAR 
EQUIPMENT 


OPERATOR  POSITION 
(CONNECTED  TO  CONTROL 

CROSSBAR  EQUIPMENT) 
INCOMfNG  TRAFFIC  SWITCHED  AT  PHILADELPHIA  TO  ANOTHER  TOLL  LINE 

FIG.  36.     Types  of  Calls  Handled  by  the  Crossbar  System  (Courtesy  Bell  System) 


to  local  subscribers,  (3)  through  calls  between  incoming  and  outgoing  dial  or  manual  toll 
lines. 

Outward  calls,  and  incoming  calls  from  toll  lines  equipped  for  toll-line  dialing,  are 
automatically  switched  under  control  of  dials  or  keysets  at  the  originating  end  of  the  lines. 


17-38 


TELEPHONY 


Incoming  calls  from  other  types  of  toll  lines  (ringdown  or  straightforward)  are  routed  to 
operator  positions. 

In  the  No.  4  system,  crossbar  switches  with  senders  and  markers  are  used  in  the  same 
general  way  as  in  the  local  crossbar  system,  with  such  variations  as  are  required  for  toll 
traffic.  The  operator  positions  which  supplement  the  mechanical  switching  system  for 
handling  terminating  and  through  calls  from  toll  lines  not  equipped  for  toll-line  dialing 
are  of  the  cordless  type,  but  cord  positions  may  also  be  used  to  facilitate  handling  calls 
over  congested  toll-line  groups.  This  system  permits  toll-line  dialing  into  a  city  having 
panel  or  crossbar  offices. 

Figure  37  shows  a  block  schematic  of  the  main  circuit  components  of  the  No.  4  crossbar 
system.  Five  types  of  senders  which  act  as  automatic  operators  are  provided,  three  for  in- 
coming and  two  for  outgoing  trunks.  For  each  incoming  call,  an  incoming  sender  is  con- 
nected, and,  unless  the  call  is  to  be  completed  over  a  manual  trunk  or  one  equipped  to 
receive  multifrequency  pulsing  at  the  distant  end  (in  which  case  only  an  incoming  sender 
is  required) ,  an  outgoing  sender  is  also  connected,  as  shown  in  Fig.  37.  When  an  outgoing 


FIG.  37.    Block  Schematic  of  Main  Circuit  Components  of  the  No.  4  Crossbar  Toll  System  (Courtesy 

Bell  System) 

sender  is  used,  the  incoming  sender  transfers,  through  the  primary  and  secondary  frames, 
all  the  digits  received  except  the  first  three,  which  are  used  by  the  marker  to  control  the 
connection  within  the  office.  Incoming  senders  are  designed  to  transmit  d-c  key  pulses 
and  outgoing  senders  to  receive  them,  the  transfer  of  digits  being  at  the  rate  of  8  per  sec. 
All  incoming  senders  are  also  arranged  to  send  out  multifrequency  pulses,  so  that  out- 
going senders  are  not  required  when  the  terminating  points  have  senders  capable  of  re- 
ceiving this  type  of  pulse,  such  as  at  local  and  toll  crossbar  offices.  Ultimately,  when 
multifrequency  pulsing  becomes  general,  outgoing  senders  may  be  eliminated.  While 
incoming  senders  are  arranged  to  send  either  d-c  or  multifrequency  pulses,  the  dial  in- 
coming sender  is  designed  to  receive  dial  pulses  (10  or  20  per  second)  recorded  on  crossbar 
switches,  and  the  key  pulsing  incoming  sender  to  receive  d-c  or  multifrequency  pulses,  as 
may  be  indicated  by  a  signal  from  the  incoming  circuit,  as  recorded  on  relays,  four  relays 
for  each  digit.  The  third  type  of  incoming  sender  (position)  is  associated  with  each  oper- 
ator's position;  on  an  outgoing  call  from  the  position,  the  sender  connects  to  a  marker, 
into  ^  which  it  passes  the  first  three  digits  received,  for  the  purpose  of  establishing  con- 
nection to  the  desired  outgoing  trunk.  When  this  trunk  is  selected  an  outgoing  sender  is 
attached,  except  when  the  trunk  is  on  the  manual  or  multifrequency  basis. 
_  Both  types  of  outgoing  senders  receive  d-c  key  pulses,  but  each  type  sends  out  different 
signals,  depending  on  the  signal  the  sender  receives  from  the  outgoing  trunk.  One  type 
receives  four  or  five  digits  and  controls  the  sending  of  either  revertive  or  call-indicator 
pulses.  Revertive  pulses  are  used  for  completing  calls  to  panel  or  crossbar  offices;  call- 
indicator  pulses,  for  calls  to  manual  offices  in  panel  areas.  The  other  type  of  outgoing 
sender  receives  up  to  11  digits,  and  either  sends  them  out  as  dial  pulses  into  a  step-by-step 


TOLL  SYSTEMS  AND  OPERATION 


17-39 


office  or  connects  itself  to  a  call-announcer  and  controls  the  sending  of  the  latter's  voice 
announcements,  which  are  limited  to  five  digits, 

Figure  38  shows  the  types  of  connections  that  incoming  and  outgoing  senders  are  re- 
quired to  control.  For  a  call  to  a  local  office  within  the  crossbar  toll  office  area,  only  the 
called  office  code  and  four  or  five  digits  are  required  to  reach  the  subscriber  from  the  cross- 
bar toll  office,  since  the  trunk  selected  by  the  crossbar  equipment  connects  directly  with 
the  called  office.  For  a  call  to  another  switching  area  through  an  intermediate  point, 
one,  two,  or  three  additional  digits  for  use  at  the  intermediate  point  must  be  dialed  or 
keyed  following  the  switching  code,  requiring  up  to  14  digits  maximum. 

All  senders  are  safeguarded  from  being  held  too  long  on  a  connection  by  timing  circuits, 
which,  after  a  predetermined  time,  signal  the  originating  operator  to  start  the  call  again, 
and  are  then  released.  When  trouble  involving  the  sender  exists,  the  sender  is  auto- 
matically held  for  inspection  and  the  maintenance  forces  are  notified  by  alarm  circuits. 


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FIG.  38.    Representative  Types  of  Calls  Switched  by  the  No.  4  Crossbar  Toll  Office  (Courtesy  Bell 

System) 

TELEPHONE  REPEATERS  are  essentially  voice-frequency  amplifying  devices  with 
suitable  talking  and  monitoring  features  designed  for  use  in  voice-frequency  toll  circuits 
which  would,  without  amplification,  be  greatly  restricted  in  their  length.  By  means  of 
such  devices,  properly  spaced  and  controlled,  toll  circuits  may  be  extended  to  any  prac- 
tical length  desired. 

In  the  early  designs,  telephone  repeaters  were  basically  of  two  types,  those  in  the  Bell 
System  being  designated  22-type  for  two-wire  circuits  and  44-type  for  four-wire  circuits. 
Present  Bell  System  practices  employ  a  single  type  of  amplifying  device  for  either  two- 
or  four-wire  circuits,  and  suitable  connecting  arrangements  to  connect  it  into  the  two-  or 
four-wire  circuits  or  any  combination  of  such  circuits. 

Figure  39  (a)  and  (&)  shows  a  schematic  diagram  of  the  two-  and  four-wire  arrange- 
ments, respectively.  "Where  a  repeater  is  employed  at  the  junction  of  a  two-  and  a  four- 
wire  circuit,  the  two-wire  arrangement  is  used  for  connection  of  the  repeater  to  the  two- 
wire  circuit  and  the  four-wire  arrangement  for  connection  of  the  repeater  to  the  four-wire 
circuit. 

In  the  two-wire  arrangement,  the  separate  branches  of  the  amplifiers  are  joined  electrically 
through  repeating  coti  hybrids  for  a  repeater  at  an  intermediate  point  in  the  telephone  cir- 
cuit. For  a  repeater  at  the  terminal  of  a  circuit  a  resistance  hybrid  arrangement  (see  Fig. 
39[c])  usually  takes  the  place  of  the  repeating  coil  hybrid  on  the  switchboard  (drop)  side 
of  the  repeater.  In  the  four-wire  arrangement  repeating  coil  hybrids  are  not  usedt  except 
where  required  at  the  junction  of  two-  and  four-wire  circuits.  Four-wire  terminating 
sets  are  employed  on  the  drop  side  of  four-wire  repeaters. 

The  repeating  coil  hybrid  arrangement  (Fig.  39(a])  consists  of  two  repeating  coils,  A 
and  B,  with  low-inductance  windings  so  related  and  connected  as  to  form,  with  associated 
equipment  and  the  line,  a  balanced  bridge  circuit  when  the  line  and  balancing  equipment 
impedances  are  equal  and  the  impedances  connected  to  terminals  2-5  of  each  coil  are  equal. 
Coils  of  different  ratios  to  match  various  line  circuit  impedances,  and  with  phantom  cir- 
cuit taps  for  securing  a  phantom  circuit,  if  desired,  are  available. 


17-40 


TELEPHONY 


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FIG.  39.     Terminating  Arrangements  used  with  Telephone  Repeaters  (Courtesy  Bell  System) 


TOLL  SYSTEMS  AND   OPERATION  17-41 

Incoming  current  from  the  line  flows  through,  the  line  windings  of  both  coils  and  induces 
equal  voltages  in  the  network  windings  9-10-11-12  of  both  coils.  Since  the  poling  of  the 
network  windings  of  coil  A  are  reversed  with  respect  to  the  network  windings  of  coil  B, 
the  resultant  voltage  across  the  network  (balancing  equipment)  is  zero.  The  voltages 
induced  in  the  2-1-6-5  windings  of  both  coils  are  likewise  equal,  and  the  power  received 
from  the  line  (deducting  coil  losses)  divides  equally  between  the  impedances  of  the  ampli- 
fier branches  connected  to  2-5  of  each  coil.  Since  these  branches  transmit  in  opposite 
directions,  the  power  received  from  the  line  from  a  given  direction  by  one  of  the  branches 
and  its  amplifier  is  effective,  while  that  received  by  the  other  branch  and  its  amplifier  is 
ineffective. 

Outgoing  current  from  the  effective  amplifier  flows  through  the  2-1-6-5  winding  of  coil 
B  and  induces  equal  voltages  in  the  line  and  network  windings  of  coil  B.  The  result- 
ing currents  in  these  windings  also  flow  through  the  corresponding  windings  of  coil  A  to 
the  line  and  balancing  equipment,  respectively.  The  currents  in  the  line  and  network 
windings  of  coil  A  induce  equal  voltages  in  the  2-1-6-5  winding  of  coil  A,  but  these  volt- 
ages are  opposing  in  phase  because  of  the  reversed  poling  of  the  network  windings,  and 
the  resultant  voltage  is  zero.  The  power  received  from  the  amplifier  (deducting  coil 
losses)  thus  divides  equally  between  the  line  and  the  balancing  equipment.  The  power 
received  by  the  line  is  useful,  while  that  absorbed  by  the  balancing  equipment  is  wasted. 

In  order  to  secure  a  satisfactory  trans-hybrid  balance,  the  windings  of  a  given  coil  must 
be  mutually  balanced  to  a  high  degree  of  precision,  but  separate  coils  need  not  be  so  highly 
balanced  with  respect  to  each  other. 

The  resistance  hybrid  arrangement  (Fig.  39[c])  is  composed  of  resistances,  condensers, 
and  the  necessary  terminating  jacks  connected  to  form  a  four-branch  balanced  lattice 
type  network.  This  network  joins  together  the  amplifier  branches  on  the  terminating 
(drop)  side  of  the  repeater  and  terminates  them  in  the  required  600-ohm  two-wire  drop. 
The  network  presents  a  600-ohm  impedance  in  all  four  directions.  The  1000-cycle  loss 
through  this  network  is  10.7  db  for  each  direction  of  transmission. 

In  the  four-wire  arrangement,  one-half  of  an  amplifier  unit  is  employed  in  each  side  of 
the  four-wire  circuit,  transmitting  in  one  direction  only.  Single  repeating  coils,  of  the 
type  used  with  two-wire  arrangements,  are  provided  to  match  the  impedances  of  the  line 
and  amplifier.  Phantom  taps  are  provided  in  these  coils,  and  the  coils  must  be  matched 
for  balance  when  phantom  circuits  are  employed. 

The  amplifier  circuit  of  the  repeater  is  shown  in  Fig.  40.  This  circuit  has  a  nominal  input 
impedance  of  600  ohms,  and  the  input  transformer  impedance  ratio  (windings  9-8  to  1-7) 
is  300  to  357,000  ohms.  The  output  transformer  has  four  windings,  of  which  winding 
9-10  is  for  the  negative  feedback  feature.  The  output  impedance  ratio  of  this  trans- 
former (windings  7-S  to  1-2)  is  21,000  to  600  ohms. 

The  amplifier  vacuum  tubes  are  heater-type  pentodes,  310  A  for  regulated  and  328  A  for 
non-regulated  battery  supply. 

Grid  bias  on  the  tubes  is  obtained  from  the  voltage  drop  in  resistances  B  and  C  and  in 
the  potentiometer  through  which  the  total  cathode  current  flows. 

The  total  gain  of  each  amplifier  is  about  35  db;  the  secondary  winding  of  the  input 
transformer  is  tapped  to  provide  a  total  gain  adjustment  of  20  db  in  4-db  steps;  and  the 
potentiometer  serves  as  an  additional  gain  control  with  a  range  of  about  5.4  db.  The 
power-carrying  capacity  of  the  amplifier  is  such  that  the  transmission  level  at  the  amp  out 
jacks  may  be  as  high  as  + 10  db  with  respect  to  the  transmitting  switchboard.  The  nom- 
inal d-c  battery  supply  is  24  volts  for  the  filamenib  current  and  130  volts  for  the  plates. 

Attenuation  equalizers  for  two-wire  repeaters  are  associated  with  the  line  equipments 
and  are  connected  to  the  amplifier  inputs  at  intermediate  repeaters  and  to  the  receiving 
amplifier  line  input  at  terminal  repeaters  (terminals  1-6  and  2-5  of  coils  A,  Fig.  39[a]). 
They  are  of  the  fixed  type  and  designed  for  repeater  sections  of  average  length.  The 
low-frequency  equalizer  for  equalization  in  the  low-frequency  range  consists  of  capacitance 
or  a  combination  of  capacitance  and  resistance,  depending  on  the  line  characteristics. 
At  two-wire  circuit  terminals,  this  equalizer  is  omitted  in  the  transmitting  side  of  the 
terminal  repeater.  High-frequency  equalization  on  two-wire  circuits  is  obtained  by  tbe 
effects  of  the  various  equalizer  units  mentioned  and  by  interaction  effects  between  the 
filter  and  the  impedances  between  which  it  is  inserted. 

Equalization  for  four-wire  circuits  (see  Fig.  39[5])  is  provided  by  a  low-frequency  equalizer 
consisting  of  a  condenser  shunted  by  a  resistance,  and  a  high-frequency  equalizer  consisting 
of  a  combination  of  resistance,  inductance,  and  capacity. 

Low-pass  filters  of  nominal  600-ohm  impedance  are  provided,  to  limit  currents  above 
the  voice  range  to  be  transmitted,  in  the  four-wire  branches  of  the  line  equipment,  as- 
sociated with  the  amplifier  input  for  the  two-wire  arrangement,  as  shown  in  Fig.  39{<z), 
Three  types  of  filters  are  provided,  having  nominal  circuit  cutoffs  of  2450,  2850,  and  3500 


17-42 


TELEPHONY 


cycles  per  second.     The  filter  is  omitted  from  the  transmitting  branch  of  terminal  re- 
peaters. 

Regulating  network  equipment  is  provided  for  insertion  in  the  repeater  circuits  (see 
Fig.  39[a]  and  [6]),  as  required,  to  compensate  for  changes  in  line  attenuation  due  to 
temperature  variations.  This  equipment  functions  under  control  of  a  pilot  wire  regulating 
system  which  actuates  relays,  causing  resistance-type  loss  pads  to  be  cut  in  or  out  of  the 
repeater  circuit,  as  required,  to  maintain  circuit  transmission  levels.  This  equipment  is 
more  fully  described  in  another  part  of  this  section. 

ODD  AMPLIFIER  CIRCUIT 


PLATE 


FIG.  40. 


EVEN   AMPLIFIER  CIRCUIT 
VI  Telephone  Repeater  Circuit  (Courtesy  Bell  System) 


The  balancing  equipment,  connected  to  9-12  of  each  A  coil  (Fig.  39[a]),  is  required  to 
balance  the  line  and  its  equipment  (up  to  the  repeating  coil  hybrid)  in  each  direction  of 
transmission  for  two-wire  arrangements  but  is  not  required  for  four-wire  arrangements, 
at  intermediate  points.  At  both  the  two-  and  four-wire  terminal  repeaters,  a  simple 
compromise  network  is  employed  on  the  drop  side  with  the  resistance  hybrid  (Fig.  39 [c]) 
and  four-wire  terminating  sets,  respectively.  Balancing  equipments  are  designed  in 
various  combinations  of  resistance,  capacitance,  and  inductance  to  closely  match  the 
impedance  of  their  various  associated  lines. 

Signaling  over  repeater-equipped  circuits  requires  the  use  of  auxiliary  signaling  circuits 
employing  20,  135,  or  1000  cycles  or  composited  d-c  signaling  on  two-wire  circuits.  The 
20-  and  135-cycle  and  composited  d-c  signals  are  by-passed  around  intermediate  re- 
peaters, through  which  such  signals  will  not  pass,  but  1000-cycle  signaling  will  pass  through 
the  repeaters  in  the  same  manner  as  voice-frequency  currents.  For  four-wire  circuits, 
in  which  there  are  usually  a  number  of  repeaters,  1000-cycle  signaling  is  generally  employed 
as  the  most  economical  and  satisfactory  arrangement. 


TOLL  SYSTEMS  AND  OPERATION 


17-43 


CARRIER  TELEPHONE  SYSTEMS  permit  the  securing  of  additional  telephone  chan- 
nels between  two  toll  centers  by  superimposing  carrier  frequencies  on  voice-frequency 
wire  circuits  between  these  points.  At  the  terminals  of  the  carrier  channels,  carrier  equip- 
ment is  required  _  which  is  capable  of  converting  voice  frequencies  to  modulated  carrier 
currents,  transmitting  them  over  the  wire  circuit,  and  reconverting  or  demodulating 
them  at  the  receiving  end  to  voice  frequencies.  This  equipment  must  operate  in  both 
directions  of  transmission.  Carrier  equipments  (carrier  repeater  or  transfer  units)  are 
also  employed  for  amplifying  carrier  currents  and  for  transferring  the  carrier  channels 
where  the  wire  circuit  does  not,  but  the  carrier  channels  do,  extend  tnxough  the  inter- 
mediate office. 

Carrier  telephone  systems  are  in  operation  in  many  countries,  but  they  are  in  use  to  the 
greatest  extent  in  the  United  States  because  of  its  vast  network  of  toll  circuits.  These 
systems  are  manufactured  by  a  number  of  different  companies,  both  Independent  and 
Bell,  the  number  of  two-way  channels  provided  in  the  various  systems  ranging  from  one 
to  twelve,  excluding  the  L-type  carrier  system.  The  Bell  System  is  the  largest  manu- 


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FIG.  41.     Channel  Frequency  Allocations  for  Telephone  Carrier  Systems  (Courtesy  Bell  System) 

facturer  and  user  of  carrier  telephone  equipment  in  this  country,  and  the  frequency  al- 
location chart  shown  in  Fig.  41  applies  to  carrier  systems  of  Western  Electric  Co.  (Bell 
System)  make.  From  this  figure,  it  is  seen  that  the  D,  G,  and  H  systems  provide  a  single 
channel,  the  C  series  three  channels  and  the  J  and  El  systems  twelve  channels,  all  two- 
way. 

In  operation,  the  basic  principle  is  the  same,  regardless  of  the  type  of  system  or  the 
number  of  channels  provided.  Figure  42  shows  a  block  schematic  of  a  type  C  carrier  tele- 
phone system  with  one  intermediate  carrier  repeater. 

It  will  be  noted  that  each  of  the  three  channels  has  identical  equipment  units  at  both 
the  east  and  west  terminals,  and  that  identical  common  equipment  is  provided  at  each 
terminal  to  serve  each  of  the  three  channels.  The  carrier  repeater  also  serves  all  three 
channels.  Voice-frequency  transmission  over  the  wire  circuit  is  not  interfered  with  by, 
nor  does  it  interfere  with,  the  carrier  currents,  because  of  the  low-pass  and  high-pass  filters 
associated  with  the  wire  line  at  each  terminal  and  at  the  repeater  or  transfer  points.  The 
low-pass  filter  will  pass  only  voice  and  the  high-pass  filter  only  carrier  frequencies. 


17-44 


TELEPHONY 


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TOLL  SYSTEMS  AND  OPERATION  17-45 

Incoming  voice  frequencies  (about  250-2750  cycles)  into  channel  1  (Pig.  42)  at  the  west 
terminal  pass  to  the  modulator,  where  they  modulate  a  carrier  frequency.  The  lower 
sideband  only  (assuming  this  system  is  of  the  CS  type)  is  transmitted  through  the  band 
filter,  transmitting  amplifier,  and  directional  and  high-pass  filters  to  the  wire  line.  At 
the  intermediate  point  the  sideband  frequencies  pass  through  the  high-pass  and  direc- 
tional filters,  equalizer,  west  to  east  amplifier,  directional  and  high-pass  filters,  thence 
to  the  line.  At  the  east  terminal  the  sideband  passes  through  the  high-pass  and  directional 
filters,  receiving  amplifier,  band  filter,  and  demodulator  to  the  voice  terminal  of  channel 
1.  Transmission  takes  place  similarly  in  the  opposite  direction.  The  band  filters  pass 
only  the  band  of  frequencies  intended  for  their  particular  channels,  blocking  out  all  fre- 
quencies of  other  channels  which  travel  the  common  paths.  The  directional  filters  sep- 
arate the  incoming  and  outgoing  bands  of  frequencies.  The  amplifiers,  which  are  of  the 
high-gain,  negative-feedback  type,  are  necessary  to  maintain  proper  levels  of  transmission 
for  the  carrier  currents.  The  pilot  equipment  indicates  and  controls  the  transmission 
levels  automatically,  so  that  manual  adjustments  are  not  required.  The  three-winding 
transformers  (hybrid  coils)  separate  the  transmitting  and  receiving  voice  paths  at  the 
voice-frequency  terminals  of  each  carrier  channel. 

For  open-wire  operation,  present  practices  make  use  of  the  G  system  for  single-channel 
very  short  circuits  (under  about  25  miles) ,  the  H  system  for  single-channel  medium-length 
circuits  (up  to  about  300  miles,  with  repeaters  about  every  125  miles) ,  and  the  C  system 
for  three-channel  groups  ranging  from  about  100  miles  to  any  length  desired  with  re- 
peaters about  every  150  miles.  The  G  system  equipment  does  not  have  an  amplifier, 
but  one  may  be  associated  with  the  system  externally.  The  CN  allocation  of  the  type 
C  system  and  the  D  system  are  not  used  for  new  installations.  All  these  systems  are  con- 
sidered to  be  in  the  low-frequency  group  of  carrier  systems  (up  to  about  30  ke),  and  they 
provide  about  2500-cycle  voice  bands. 

The  broad-band  carrier  systems  operate  in  a  range  from  about  12  to  2000  kc  or  more  and 
provide  about  3000-cycle  voice  bands.  As  may  be  noted  in  Fig.  41,  the  K  system,  for 
toll  cable  use,  functions  between  12  and  60  kc;  the  J  system,  for  open  wire,  between  36 
and  140  kc;  and  the  L  system  (not  shown),  for  coaxial  cable  use  only,  between  60  and  2000 
or  more  kc.  The  J  and  K  systems  may  be  operated  any  distance  desired  but  require  re- 
peaters about  every  70  and  16  miles  respectively. 

The  L  type  carrier  system,  operating  over  specially  designed  conductors,  known  as 
coaxial  cable,  because  of  their  construction,  is  capable  of  providing  up  to  480  two-way 
circuits  per  pair  of  coaxials  (depending  on  the  make-up  of  the  cable)  if  coaxial  amplifiers 
are  spaced  about  5  to  8  miles  apart.  Practically,  however,  the  number  of  circuits  to  be 
provided  in  any  L  system  will  depend  upon  traffic  requirements,  since  circuits  can  be  added 
as  desired  at  any  time  in  groups  of  12. 

Figure  43  shows  the  frequency  translations  which  take  place  at  an  L  type  carrier  tele- 
phone system  terminal.  Three  steps  of  modulation  are  employed  to  change  an  individual 
voice-frequency  channel  of  0  to  4  kc  to  its  proper  line  frequency  assignment.  The  first 
step  of  modulation  occurs  in  the  channel  bank  and  translates  a  group  of  12  voice  channels 
to  the  60  to  108  kc  frequency  band.  The  second  step  of  modulation  occurs  in  the  group 
modulators  and  moves  each  fundamental  group  of  12  channels  each  (60  to  108  kc)  to  one 
of  five  frequency  assignments  (each  48  kc  wide)  within  the  312  to  552  ke  band.  This 
band,  designated  a  basic  supergroup,  is  240  kc  wide  and  accommodates  60  channels  of  4 
kc  each.  The  third  step  of  modulation  occurs  in  the  supergroup  modulators  and  translates 
each  basic  supergroup  of  60  channels  each  (312  to  552  ke)  to  one  of  eight  frequency  as- 
signments (each  240  kc  wide)  within  the  68  to  2044  kc  band.  The  480  voice-frequency 
channels  take  line  frequency  positions  within  the  68  to  2044  kc  band,  and  the  four  pilot 
channels  are  assigned  to  frequencies  of  64,  556,  2064,  and  3096  kc.  The  supergroups  are 
separated  by  8  kc  each,  except  that  4  kc  and  12  kc  separate  the  first  and  second  supergroups 
and  the  second  and  third  supergroups,  respectively. 

All  long-haul  carrier  systems,  including  the  C,  JT  K  and  L  systems,  have  regulating  and 
pilot  channel  equipment  which  automatically  adds  transmission  gain  or  loss  as  is  neces- 
sary to  maintain  the  channels  within  predetermined  transmission  equivalents.  For 
cable  facilities  the  normal  transmission  changes  are  due  to  temperature  variations,  being 
greater  in  aerial  than  in  underground  cable.  Flat  gain  regulators  with  master  controllers 
are  employed  in  every  repeater  section  for  K  systems.  However,  the  amount  of  at- 
tenuation variation  in  cable  pairs  with  a  given  change  in  temperature  is  not  the  same  for 
all  frequencies,  an  effect  known  as  twist.  To  overcome  twist  effects,  correcting  circuits 
are  also  provided  about  every  100  miles  for  aerial  and  200  miles  for  underground  cable. 

Large  economies  are  possible  with  carrier  systems,  principally  to  provide  additional 
toll  circuits  between  toll  centers.  Their  usage  avoids,  in  many  cases,  the  stringing  of  ex- 


17-46 


TELEPHONY 


pensive  wire  circuits  or  possibly  building  a  new  pole  line  for  open  wire  or  cable  or  placing 
underground  cable.     In  any  event,  studies  will  indicate  the  economies  involved. 

The  basic  12-channel  equipment  units  of  the  J,  K,  and  L  systems  are  similar,  the  channel 
modulators  elevating  the  voice-frequency  bands  for  12  channels  (4  kc  per  channel)  from 
0-48  kc  up  to  60-108  kc  for  these  systems.  Similarly  the  channel  demodulator  receives 
the  12  channels  at  60  to  108  kc.  The  K  system  operates  over  separate  toll  cable  pairs  in 
separate  cables  with  carrier  repeaters,  using  the  12  to  60  kc  band  in  each  direction  in  the 

CARRIER 
FREQUENCIES 

OF  — *3096  PILOT 

SUPERGROUP  -»>2064  PILOT 


MOTES: 

U-  CHANNELS  ARE  UPPER 

SIDEBANDS 
L-  CHANNELS  ARE  LOWER 

SIDEBANDS 

FREQUENCIES  ARE  IN  KILO- 
CYCLES PER  SECOND 


MODULATORS   |-p 


JlJ  1556 


BASIC 
SUPERGROUP 

OF  60 
CHANNELS, 


CARRIER      I 
FREQUENCIES  \ 
OF  GROUP     { 
.MODULATORS,! 


FUNDAMENTAL 
GROUP  OF  12 
CHANNELS 

LOWER 
.SIDEBANDS    , 

Y 

12  CIRCUITS  * 


612  -» 
564-» 
516  -» 
468-* 
420  -»• 

_5_ 

~Z- 

\ 

k 

552 
504 
456  1 
408f 
360 
312J 

2044 


1804 
1796 


T|  1052 


-5j  812      § 


804 


8E 


-    556  PILOT 


LU312 


— *•  64  PILOT 


40  OF  THESE  MODULATED  I 

IN  GROUPS  OF  5  MAKING   8  OF  THESE 


FIG.  43.     Frequency  Translations  at  an  L-type  Carrier  Telephone  System  Terminal  (Courtesy  Bell 

System) 

cables.  The  T  system  employs  the  same  open-wire  circuit  in  both  directions,  using  line 
frequencies  of  36  to  84  kc  west  to  east  and  92  to  140  kc  east  to  west.  The  L  system  em- 
ploys two  coaxial  units  for  the  two  directions  of  transmission,  hence  the  same  frequency 
band,  from  60  to  the  required  top  kilocycle  frequency,  in  each  direction.  Group  frequency 
modulators  move  12-ehannel  groups  from  one  frequency  range  to  another,  as  required. 

Figure  44  shows  a  block  schematic  of  a  type  J  carrier  telephone  terminal;  Fig.  45,  a 
type  K  carrier  telephone  terminal  in  more  detail;  and  Fig.  46  a,  frequency  translation  di- 
agram for  the  type  J  system. 

The  particular  range  of  frequencies,  60  to  108  kc,  to  which  the  voice  frequencies  are  first 
elevated  in  the  J,  K,  and  L  systems  was  chosen,  because  (1)  high-grade  crystal  niters  can 
be  most  economically  built  for  operation  in  this  general  range,  (2)  the  second  harmonic 
(120  ke>  of  the  lowest  frequency  (60  kc)  lies  well  above  the  highest  frequency  (108  kc), 
precluding  the  possibility  of  interference  between  the  second  harmonic  (from  any  channel)' 
and  other  channels,  and  (3)  manufacturing  economies  are  achieved  by  using  the  same 


TOLL   SYSTEMS   AND   OPERATION 


17-47 


17-48 


TELEPHONY 


13S  ^V^JIWH31 


TANDEM  SYSTEMS  AND  OPERATION 


17-49 


500 


.      **-^   2 

-i  i  -I  s 


WEST- EAST 

FIG.  46.     Frequency  Translations  in  Type  3  Carrier  Systems  (Courtesy  Bell  System) 

group  of  channel  carrier  frequencies  for  all  the  broad-band  systems.  The  line  frequencies 
12  to  60  kc  were  chosen  for  the  K  system  because,  for  non-loaded  cable  pairs  Goading  not 
being  available  for  these  frequencies),  the  attenuation  increases  with  frequency  and  in  the 
frequency  range  chosen  the  attenuation-frequency  increase  is  about  uniform. 


4.  TANDEM  SYSTEMS  AND  OPERATION 

Local  tandem  offices  are  provided,  usually  one  to  each  extensive  multioffice  area,  where 
a  large  number  of  offices  are  located  within  and  adjacent  to  city  boundaries.  The  tandem 
office  is  generally  centrally  located  to  the  offices  it  serves  and  has  direct  trunks  to  all  these 
offices  and  also  to  attended  PBX  (private  branch  exchange)  switchboards  having  a  number 
of  pay  stations,  to  the  toll  board,  and  to  various  special  service  boards.  It  is  neither 
economical  nor  practical  in  large  metropolitan  centers  to  provide  a  group  of  outgoing  and 
a  group  of  incoming  direct  trunks  between  each  pair  of  offices  in  the  center,  since  cable 
plant  and  the  terminating  trunk  equipments  required  would  be  too  costly.  For  many 
combinations  of  two  offices  some  distance  apart  the  traffic  is  usually  so  light  as  to  preclude 
the  use  of  expensive  cable  pairs  between  them  for  trunk  purposes.  Thus,  the  tandem  office 
provides  an  economical  means  of  trunknig  calls  in  the  larger  centers  because  a  common 
trunk  group  is  provided  between  each  office  served  and  tandem, 

Although  the  introduction  of  a  tandem  office  in  the  call  trunHng  system  of  a  given  area 
adds  another  office  to  the  system  and  may  materially  lengthen  the  trunk  mileage  between 
certain  pairs  of  offices,  depending  on  the  geographical  location  of  the  tandem  office  with 
respect  to  the  surrounding  offices,  it  is  evident  that  two  groups  of  trunks  (one  outgoing  and 
one  incoming)  between  each  pair  of  offices  in  a  large  center  of,  say,  50  offices  would  result 
in  an  uneconomical  trunk  network  and  equipment  layout  and  a  more  complex  arrangement 
of  handling  traffic.  Also,  the  greater  number  of  calls  handled  by  the  tandem  trunk  groups 
increases  their  efficiency  and  may  largely  offset  the  tandem  layout  costs.  Tandem  offices 
are  warranted,  however,  in  any  particular  case  only  if  a  comprehensive  study  of  all  the 
various  factors  involved  indicates  their  need. 

One  system  of  tandem  trunking  now  in  use  provides  for  trunMng  from  a  local  manual 
office  to  a  panel  sender  tandem  office  on  a  straightforward  basis.  The  local  A  operator, 
upon  receiving  a  call,  inserts  the  calling  plug  of  an  A  cord  circuit  into  an  idle  outgoing 
trunk  to  tandem,  causing  selection  to  be  made  at  the  tandem  office  of  an  idle  operator  and 


17-50 


TELEPHONY 


a  tone  to  be  sent  back  to  the  A  operator  indicating  that  the  tandem  operator  is  ready  to 
receive  the  call.  The  A  operator  passes  the  called  office  name  and  number  over  the  trunk 
to  the  tandem  operator,  who  registers  this  information  on  a  keyset  (a  unit  composed  of 
individual  keys  having  office  code  and  number  designations),  with  which  each  tandem 
position  is  equipped.  The  operation  of  the  keys  in  the  keyset  causes  pulses  or  number 
announcements  to  be  sent  out  over  the  tandem  to  called  office  trunk,  thus  reaching  the 
called  office. 

If  the  called  office  is  of  the  step-by-step,  panel,  or  crossbar  type,  the  incoming  pulses  from 
tandem  reach  selector  (in  step-by-step)  or  sender  equipment  (panel  or  crossbar)  in  the 
called  office,  which  completes  the  connection  to  the  called  subscriber  in  the  regular  man- 
ner, as  previously  described  for  these  systems. 

If  the  called  office  is  of  the  manual  type,  the  incoming  pulses  reach  call  announcer 
equipment  at  the  tandem  office  before  passing  out  over  the  trunk  where  they  are  con- 
verted from  pulses  to  spoken  numbers  which  reach  an  idle  3  operator's  ear  at  the  called 
office;  or,  where  call  announcer  equipment  is  not  provided,  the  tandem  sender  equipment 
sends  out  pulses  over  the  trunk  to  call  indicator  equipment  at  the  called  office,  which  causes 
the  called  number  to  be  'displayed  before  the  B  operator.  In  either  case  the  B  operator 
connects  the  plug-ended  trunk  circuit  over  which  the  call  is  being  routed  into  the  called 
subscriber  multiple  jack,  and  ringing  automatically  starts  on  the  called  line.  Supervision 
of  the  call  at  the  local,  tandem,  and  B  boards  is  by  means  of  the  usual  lamp  signals. 

If  the  calling  subscriber  is  in  a  mechanical  office  (step-by-step,  panel,  or  crossbar),  the 
call  is  usually  routed  to  a  special  board  in  the  calling  office.  The  special  operator  passes 
the  call  to  tandem  on  a  straightforward  basis,  and  the  connection  is  then  completed  by  the 
tandem  operator  in  the  same  manner  as  described  for  a  calling  local  manual  office. 

Crossbar  tandem  equipment  is  now  standard  for  new  tandem  offices,  rather  than  the 
manual  tandem  arrangement,  just  described,  where  it  is  applicable.  This  equipment 
handles  calls  over  two-wire  trunks  from  panel  or  crossbar  offices  to  other  panel,  crossbar, 
or  panel  indicator  manual  offices  by  means  of  crossbar  switches  in  a  marker  system  of 
operation.  Calls  from  a  manual  office  through  crossbar  tandem  would  reach  a  tandem 
operator  over  straightforward  trunks  and  be  completed  as  described  for  the  panel  sender 
tandem  office. 

The  major  switching  frames  in  crossbar  tandem  offices  correspond  somewhat  to  the 
frames  in  a  local  crossbar  office.  Incoming  trunks,  terminated  on  incoming  trunk  frames, 
connect  through  trunk  link  frames,  an  office  junctor  grouping  frame,  and  office  link  frames 
to  outgoing  trunks,  which  terminate  in  incoming  trunk  equipment  in  other  offices  as  shown 
in  Fig.  47. 


TERMINATING 
OFFICE 


ORIGINATING 
OFFICE 


T,R  &  SI 


T,  R  &  S1 


T  &  R      PANEL  TANDEM  USING 
..  TRUNKS  IN  COMMON 

WITH  CROSSBAR  TANDEM 


MAIN 

DISTRIBUTING 
FRAME 


FIG.  47.     Crossbar  Tandem  Office  Units — Crossbar  System  (Courtesy  Bell  System) 


When  a  ^call  reaches  the  tandem  office  incoming  trunk  equipment,  tandem  sender  and 
marker  units  are  caused  to  associate  with  the  trunk  and  li^Tr  frames  involved  in  the  call. 
The  sender  receives  and  registers  the  incoming  pulses  from  the  originating  sender  on  a 
revertive  (pulses  sent  back)  or  dial  pulse  basis,  and  it  controls  the  tandem  switch  selec- 
tions and  the  selections  in  the  called  office.  The  marker  which  is  associated  with  the 
sender  through  the  marker  connector  frame  routes  the  call  through  the  tandem  equip- 
ment under  control  of  the  tandem  sender  registrations. 


COMMON  SYSTEMS  17-51 

5.  AUXILIARY  SERVICE  EQUIPMENT 

SERVICE  OBSERVING  DESKS  are  provided,  where  required,  at  individual  offices  or 
at  a  central  location  for  observing  the  performance  of  switchboard  operators  and  switch- 
ing equipment. 

One  type  of  service  observing  desk  is  intended  primarily  as  a  non-centralized  observing 
bureau  for  use  with  toll  or  local  crossbar,  step-by-step  and  manual,  or  combinations  of  toll 
and  local  plant.  This  desk  is  generally  employed  where  not  more  than  one  desk  position 
is  required,  and  observations  are  confined  to  circuits  at  the  same  location  as  the  desk. 
This  desk  is  equipped  with  a  cord  circuit,  operator's  telephone  circuit,  jacks,  lamps,  keys, 
and  other  equipment  for  observing  on  lines  and  trunks. 

For  mechanically  operated  systems,  pen  registering  equipment  is  also  employed  to  re- 
cord the  subscriber  dial  and  line  registering  pulses,  as  required. 

Central  observing  desks  require  direct  lines  to  the  various  offices  which  are  to  be  ob- 
served^and  as  many  desk  positions  as  necessary  to  make  the  desired  observations. 

The  information  obtained  by  means  of  these  desks  is  a  very  important  aid  in  improving 
operator  and  equipment  efficiency  and  in  bettering  service  generally  to  the  public. 

INFORMATION  DESKS  are  provided  at  individual  offices  or  central  locations  to 
furnish  subscribers  with  information  about  telephone  numbers  not  listed  in  the  telephone 
directory  or  changed  from  the  directory  listing  and  about  many  other  items  essential  in 
assisting  the  subscribers  to  secure  a  wholly  satisfactory  telephone  service.  Up-to-date 
files  of  telephone  listings  are  maintained  at  the  information  desks  accessible  to  each  in- 
formation operator.  Centralized  information  desks  require  that  subscribers  in  each 
office  be  routed  over  the  trunMng  system,  provided  for  such  service,  to  the  centralized 
desk.  Individual  office  information  desks  are  reached  over  direct  intraoffice  trunks. 

CHIEF  OPERATOR  DESKS  are,  if  required,  located  one  in  each  office.  Various  types 
of  calls  are  referred  to  the  chief  operator,  who  is  in  direct  charge  of  the  operating  forces. 
Complaints  from,  the  subscriber  about  the  service  rendered  or  any  other  items  regarding 
the  wishes  of  the  subscriber  in  connection  with  calls  the  handling  of  which  is  not  the 
operator's  function  are  referred  to  the  chief  operator  or  a  supervisor. 
''•  Trunks  are  provided  from  manual  or  DSA  boards  to  the  chief  operator  desks,  over  which 
these  calls  are  routed. 

REPAIR  SERVICE  DESKS  are  required  in  all  but  the  very  small  offices  to  receive 
subscriber  complaints  of  service  or  reports  of  trouble  encountered  with  the  substation 
equipment  or  telephone  plant  in  general.  Many  other  types  of  reports  from  the  public 
are  also  referred  to  these  desks,  which  are  convenient  contact  points  for  subscribers.  Rec- 
ords are  maintained  of  troubles  reported  and  cleared  on  subscriber  lines,  and  valuable  data 
are  secured  from  them  for  studies  of  troubles  and  their  elimination. 

Repair  trunks  are  provided  at  individual  offices  or  to  a  central  point  for  receiving  these 
calls  except  in  small  offices  where  such  calls  are  usually  received  at  local  testboards. 

6.  COMMON  SYSTEMS 

MAIN  DISTRIBUTING  FRAMES  (MDF)  are  required  in  central  offices  for  terminat- 
ing the  outside  local  and  toll  lines,  which  are  usually  brought  into  the  office  on  cable  pairs. 
These  pairs  terminate  directly  on  protector  strips,  mounted  vertically  on  the  vertical  side 
(VMDF)  of  the  frame.  In  the  smaller  offices  local  and  toll  MDF  are  usually  combined; 
in  the  larger  installations,  involving  a  number  of  toll  and  toll  entrance  cables,  separate 
local  and  toll  frames  are  provided.  Terminal  strips  with  insulated  ,metal  terminals  are 
mounted  horizontally  in  rows  (shelves)  on  the  horizontal  side  (HMDF)  of  the  frame,  and 
these  terminals  are  cabled  to  an  intermediate  distributing  frame  (IDF).  Cross-connecting 
wire  (Jumpers)  may  be  run  between  any  vertical  protector  strip  and  any  horizontal  ter- 
minal strip  so  that  any  incoming  line  on  the  VMDF  may  be  connected  to  any  pair  of  ter- 
minals on  the  HMDF.  Other  arrangements  of  protector  and  terminal  strips  are  also 
used,  depending  on  the  needs  of  the  individual  office. 

INTERMEDIATE  DISTRIBUTING  FRAMES  (IDF)  are  usually  employed  in  both 
manual  and  mechanical  offices  for  local  and  toll  lines.  These  frames  have  terminal  strips 
on  both  the  horizontal  (HIDF)  and  the  vertical  sides  (VTDF)  of  the  frame.  For  manual 
local  lines  cabling  from  the  MDF  is  terminated  on  the  HIDF  terminal  strips,  to  which 
cahling  from  the  manual  A  and  B  boards  is  also  multipled.  For  small  subscriber  lamp 
multiple  and  single  office  boards,  the  B  board  and  its  cabling  are  omitted.  The  subscriber 
answering  jacks  in  the  A  board  and  the  line  circuit  equipment  are  cabled  to  the  vertical 
terminal  strips  (VIDF) .  Cross-connecting  wire  may  be  run  between  any  horizontal  and 


17-52 


TELEPHONY 


GROUNDED 


any  vertical  terminal  strip,  thus  providing  a  means  of  connecting  any  outside  line  to  any 
subscriber  A.  and  B  board  multiple  jack  and  to  any  answering  jack  and  line  circuit.  By 
properly  distributing  heavy  and  light  calling  lines  throughout  the  switchboard,  traffic  loads 
can  be  more  uniformly  spread  over  the  operator  positions. 

Relay  racks  are  also  provided  for  mounting  various  types  of  apparatus,  such  as  relays, 
repeating  coils,  apparatus  mounted  on  panels,  testing  equipment,  and  many  other  equip- 
ment units. 

PROTECTORS  are  provided  at  the  vertical  side  of  the  MDF  in  all  central  offices  having 
exposed  outside  cable  or  open  wire  plant  in  order  to  protect  the  equipment  from  damage 
due  to  excessive  voltages  and  currents  from  foreign  sources  such  as  lightning  and  power 
lines.  Figure  48  shows  a  typical  protector  used  at  main  frames.  Where  the  outside  cable 
enters  aerially  and  is  exposed  to  these  foreign  sources,  as  it  may  be  in  small  offices,  fuses 

are  also  required  in  the  circuits,  unless  the 

I  CABLE   CONDUCTORS    enterm£  Cable  haS  at  leaSt  6  f  *  °f  ^  Or  finer 

\  TO  OUTSIDE  PLANT  gaSe  cable  inserted  in  it  in  such  manner  that 
no  power  line  contacts  can  occur  between 
the  point  of  insertion  and  the  MDF. 

The  protector  consists  principally  of  a 
spring  assembly,  arranged  to  hold  two  heat 
coils  and  two  sets  of  protector  blocks,  one  of 
each  for  each  side  of  a  metallic  line.  The 
heat  coils  designed  to  protect  delicate  cen- 
tral office  equipment  usually  operate  if  0.35 
amp  flows  for  3  hours  or  longer  or  if  0.54 
amp  flows  for  more  than  210  sec.  This  coil 
consists  of  a  small  coil  of  wire  wound  around 

JL        IJI-Jl—  •  HEAT  coiu  a  c°PPer  tuke»  into  which  is  inserted  a  metal 

I          all   T!"  pm  ^e^  *n  p*ace  ky  easily  melting  solder. 

I          jlBfl  ^ke  c°il  °f  wi16  *s  placed  in  series  with  the 

wy  1  vW^K  jjne  conductor,  and  if  sufficiently  heated  the 

solder  melts,  releasing  the  metal  pin,  which 
is  forced  against  a  grounding  spring  by  the 
outside  line  spring  of  the  protector.  Several 
types  of  protectors  and  heat  coils  are  avail- 
able; some  of  the  heat  coils  open  the  line 
conductor  when  they  operate  and  may  be 


TO  CENTRAL 
OFFICE  EQUIPMENT 


PROTECTOR  BLOCKS 


i_J  -------  CARBON 

L  -----  L  -------  PORCELAIN 


FIG.  48.     Main  Frame  Protector  with  Heat  Coils 
and  Protector  Blocks  (Courtesy  Bell  System) 


reset  mechanically  without  replacing  or 
resoldering  the  coil.  The  protector  blocks 
consist  of  a  porcelain  block  with  carbon  in- 
sert and  a  solid  carbon  block,  a  pah-  to  each  side  of  the  line,  the  air  gap  between  the  insert 
and  carbon  block  being  about  0.003  in.  This  gap  will,  on  the  average,  break  down  at 
about  350  volts  potential. 

The  protector  ground  bars  are  bonded  to  a  main-frame  ground  bar  which  is  connected 
to  the  office  ground. 

TESTBOARDS  are  provided  in  all  attended  central  offices,  and  testing  equipment  of 
suitable  design  in  all  offices,  for  the  purpose  of  deterrnining  the  condition  of  lines  and 
equipment,  either  because  of  reported  or  indicated  trouble  or  as  a  trouble-preventive 
measure. 

Many  types  of  test  cabinets  and  testboards  have  been  made  available  to  meet  the  needs 
of  the  particular  location  in  which  they  may  be  installed. 

In  small  manual  local  offices  test  cabinets,  equipped  with  voltmeter  (with  batteries), 
test  keys  and  cords',  trunks,  and  in  some  cases  an  external  Wheatstone  bridge,  are  em- 
ployed. In  large  manual  installations  a  number  of  positions  of  testboard  may  be  used, 
depending  on  the  volume  of  repair  and  subscriber  installation  work  involved.  Each  of 
these  positions  usually  has  a  voltmeter,  test  keys  and  cords,  trunks,  and  access  to  one 
Wheatstone  bridge,  as  required.  Lines  and  equipment  are  tested  on  reports  of  trouble 
or  after  installation  work  is  completed,  and,  in  case  of  trouble,  its  location  is  determined 
and  the  trouble  is  cleared  by  outside  repairmen. 

In  manual  toll  offices  testboards  (primary,  secondary,  or  both)  are  arranged  with  a 
group  of  jacks  for  each  toll  circuit,  which  is  wired  through  these  jacks  before  reaching  the 
toll  switchboard.  The  test  positions  are  provided  with  voltmeter,  Wheatstone  bridge, 
test  keys  and  cords,  and  trunks  for  testing  lines  and  equipment  and  locating  and 
clearing  troubles.  The  testboard  jacks  also  serve  to  patch  lines  and  equipment  to  spare 
lines  and  equipment  when  the  regular  layout  is  in  trouble  or  to  make  temporary  circuit 
changes. 


POWER  SYSTEMS  17-53 

In  step-by-step  local  or  toll  offices  the  testboard  equipment  is  comparable  to  that  provided 
in  manual  offices  except  that  the  positions  are  equipped  with  rtmls  and  dialing  trunks  for 
connecting  to  the  various  lines  and  equipments. 

In  panel  died  and  crossbar  local  offices  a  local  test  desk  test  selector  frame  is  employed, 
consisting  of  a  bay  for  sequence  switches  and  a  bay  for  relays,  with  terminal  strips  located 
at  the  side  of  the  frame.  This  frame  is  used  to  associate  the  local  test  desk  with  the  final 
selectors  which  are  arranged  for  testing  subscriber  lines.  The  operation  of  the  cutoff 
relay  in  the  subscriber  line  circuit  may  be  controlled  from  a  key  in  the  test  desk.  The 
six  first  selector  circuits  which  connect  to  trunk  circuits  in  the  test  desk  have  access  to  any 
one  of  15  second  selectors,  which  terminate  in  final  test  selectors.  Test  trunk  ringing 
circuits  are  furnished  as  required.  The  test  desk  is  equipped  with  voltmeter,  test  keys 
and  cords,  trunks,  Wheatstone  bridge  if  desired,  and  other  necessary  apparatus  for  check- 
ing the  condition  of  and  locating  and  clearing  trouble  on  lines  and  equipment.  In  addition 
to  the  test  desk  there  are  various  frames  for  testing  the  operating  condition  of  the  equip- 
ment, such  as  the  outgoing  trunk  test  frame,  decoder  test  frame,  and  multiple  registra- 
tion test  unit. 

In  crossbar  toll  offices  a  toll  test  board  is  provided  for  making  overall  tests  on  the  toll 
trunks  in  order  to  locate  troubles  and  restore  the  trunks  to  normal.  A  jack  field  is  pro- 
vided through  which  the  intertoll  trunks  are  wired,  and  miscellaneous  trunks  are  ter- 
minated in  this  field.  The  circuits  in  this  testboard  are  four-wire,  requiring  twin  plugs. 
For  talking  and  monitoring,  the  circuits  are  reduced  to  two-wire  by  means  of  a  hybrid 
coil.  A  transmission-measuring  system  with  the  readings  projected  on  screens  at  the 
ends  of  the  testboard,  a  noise-measuring  unit,  and  a  variable  oscillator  are  provided  as 
part  of  the  testing  equipment.  Outgoing  toll  trunks  may  be  locked  out  (made  busy)  and 
tested  by  dialing  through  the  trouble  tracing  frame,  which  seizes  the  incoming  trunk  con- 
nected to  the  outgoing  trunk  in  trouble  or  to  be  tested  and  lights  a  lamp  associated  with 
that  trunk  at  the  testboard.  The  desired  outgoing  trunk  may  also  be  seized  by  plugging 
a  test  cord  into  the  test  jack  appearance  of  the  incoming  intertoll  trunk  and  operating  the 
lockout  relay  in  the  outgoing  trunk  through  the  connecting  switches. 

In  addition  to  the  toll  testboard  a  maintenance  center  is  provided  for  each  No.  4  crossbar 
toll  system,  in  which  various  testing  frames  are  located  with  a  chief  switchman's  desk  and 
files  in  front  of  the  frames. 

The  maintenance  forces  at  this  center  are  occupied  with  responding  to  alarms,  making 
tests,  following  up  trouble  reports  and  assistance  requests,  and  maintaining  records  of  the 
operations. 

7.  POWER  SYSTEMS 

Power  equipment  for  central  offices  is  required  to  provide  direct  current  for  talking  and 
signaling  and  alternating  and  pulsating  current  for  signaling  and  for  many  other  aux- 
iliary needs  in  telephone  operations. 

Power  equipment  for  magneto  offices  usually  consists  of  dry  cells  or  a  battery 
eliminator  to  supply  direct  current  to  the  operator's  telephone  set  and  a  hand  generator 
and  power-operated  ringing  device  for  ringing  subscriber  bells.  Magneto  subscriber 
telephones  are  supplied  transmitter  current  by  dry  cells  at  each  telephone  or  other  power 
sources. 

Power  equipment  consists  of  motor-generator  sets  and  rectifier  units  (in  common-battery 
offices)  for  supplying  direct  current  for  the  energization  of  subscriber  telephone  trans- 
mitters, private  branch  exchanges  or  private  automatic  exchanges,  central  office  cord  and 
operators'  circuits,  relays,  switches,  alarms,  carrier  systems,  telephone  repeaters,  and 
many  other  central-office  units.  Continuity  of  service  is  insured  by  the  provision  of 
storage  batteries  which  float  across  the  d-c  office  power  supply  and,  in  case  of  commercial 
power  failure,  will  carry  the  office  load  for  a  short  period  of  time,  and  by  duplicate  charging 
units  operated  by  other  than  commercial  electric  power. 

The  types  of  charging  and  load  supplying  power  units  (including  storage  batteries)  and 
of  the  signal  supply  units  vary  over  a  wide  range  of  equipments,  which  have  different 
capacities,  functions,  and  characteristics  depending  on  the  office  power  load  demands  and 
the  purposes  for  which  the  units  are  designed. 

Att  common-battery  offices,  both  local  and  toll,  require  24-  and  48-volt  d-c  supply,  which 
may  be  provided  by  regulated  rectifiers  (mercury  vapor,  tungar,  copper  oxide,  or  selenium 
types)  for  the  smaller  loads;  and  motor-generator  sets,  singly  or  in  multiple  units,  for  the 
larger  loads.  In  all  cases,  storage  batteries  of  suitable  capacity  to  provide  for  the  24-  and 
48-volt  demand  are  bridged  across  the  d-c  power  supply  leads  on  a  float  basis;  that  is,  the 
office  load  is  carried  by  the  generating  units,  the  batteries  serving  as  an  emergency  source 
of  power.  These  offices  may  also  require  a  130-volt  plate  supply  for  carrier  systems, 


17-54 


TELEPHONY 


telephone  repeaters,  and  other  vacuum-tube  devices,  which  supply  may  be  furnished 
from  regulated  rectifiers  for  small  loads  or  from  motor-generator  sets  for  the  larger  loads. 
Storage  batteries  of  the  proper  capacity  are  also  required  across  this  supply  on  a  float 
basis  for  emergency  reasons.  Figure  49  shows  a  typical  central-office  power-plant  ar- 
rangement with  a  single  floating  battery  and  automatic  control.  Although  only  one 
motor-generator  set  is  shown,  it  is  the  usual  practice  to  add  these  units  for  multiple  opera- 
tion as  required. 

Ringing  and  signal  supply  units  may  be  of  the  vibrator  or  subcycle  converter  type  for 
small  offices  or  motor-generator  sets  for  the  larger  offices.  These  units  are  designed  to 
supply  the  usual  20-cycle,  75  to  175  volt  ringing  power,  but  certain  multifrequency  sets 


EMERGENCY   CELL 

SWITCH 
CONTROL 


EMERGENCY^"!"  J**         .JZ, 


CELLS 
COUNTER 


MOTOR-GENERATOR 
SET 


1          >     I 
III, 

TO  SECOND 

MOTOR- GENERATOR 

SET 

FIG.  49.    Typical  Central  Office  Power  Plant  Arrangement  with  Single  Floating  Battery  and  Auto- 
matic Control  (Courtesy  Bell  System) 

are  also  available  for  supplying  harmonic  frequencies  of  16  2/3,  33  1/s,  50,  66  2/s,  and,  if 
desired,  25  cycles,  at  from  75  to  175  volts,  for  selective  party-line  ringing.  These  units 
also  provide  110  to  120  volt  d-c  coin  control  supply,  superimposed  ringing  with  46  to  52 
volt  silent  interval  tripping  battery  supply,  howler  tone  (applied  to  lines  with  receivers 
left  off  the  hook),  interrupter  tones  of  various  frequencies  for  operator  and  subscriber 
signals,  such  as  busy,  trouble,  or  operating  procedure. 

In  addition  to  the  above  types  of  signaling  units  there  are  several  different  types  of 
single  and  multifrequency  generators  producing  frequencies  for  signaling,  such  as  135  and 
1000  cycles,  and  for  vacuum-tube  oscillators. 

All  power  equipment  is  automatically  regulated  to  close  limits. 


RADIO  TELEPHONE  SYSTEMS 


Radio  telephone  systems  have  been  in  operation  for  over  25  years,  but  with  the  growing 
need  for  such  systems  in  recent  years,  accelerated  by  World  War  II,  they  are  rapidly 
coming  into  use  for  many  purposes  in  the  communications  field.  Many  of  these  systems 
are  arranged  for  connection  to  wire  telephone  plant,  so  that  subscriber  stations  may  be 
interconnected  over  radio  channels  and  wire  line  extensions  to  give  local,  national,  and 
world-wide  service. 


APPLICATIONS 


17-55 


8.  APPLICATIONS 

Radio  telephone  service  is  being  employed  today  (1)  between  all  of  the  larger  and  many 
small  countries,  (2)  within  individual  countries,  (3)  between  ships  at  sea  and  fixed  land 
stations,  (4)  between  coastal  harbor  and  inland  waterways  ships  and  fixed  land  stations, 

(5)  between  mobile  vehicles  of  various  classes  and  stationary  points  or  other  mobile  units, 

(6)  between  planes  and  ground  stations  or  other  flying  planes,  (7)  directly  between  persons 
(walkie-talkie),  and  (8)  for  special  and  emergency  use,  such  as  for  fire,  police,  emergency 
repair  units,  and  for  temporarily  bridging  gaps  in  telephone  lines  that  have  sustained 
major  damage. 

Development  work  and  trial  tests  are  in  progress  for  the  use  of  radio  channels  (1)  be- 
tween and  within  trains  and  railroad  operating  control  points,  and  (2)  for  rural  line  tele- 
phone service.  An  important  activity,  now  in  the  experimental  stage,  is  the  use  of  super 
high-frequency  (microwave)  radio  systems  for  toll  telephone,  television,  facsimile,  and 
other  services.  Undoubtedly  many  other  applications  of  value  to  the  public  will  be  found 
for  radio  telephone  systems. 

Radio  telephone  systems  may  be  listed  at  present  under  the  following  general  classi- 
fications: 


1.  Long-haul  toll. 

2.  Short-haul  toll. 

3.  Coastal  harbor  and 
inland  waterways. 

4.  Highway  mobile. 


5.  Urban  mobile. 

6.  Airways  mobile. 

7.  Railway  mobile. 

8.  Rural  subscriber. 

9.  Special  emergency. 


Table  1  lists  some  of  the  principal  operating  data  regarding  these  systems. 

A  new  microwave  radio  relay  system,  using  frequencies  between  about  2000  to  12,000  or 
more  megacycles,  is  installed  between  New  York  and  Boston.  This  system,  using  "line- 
sight"  frequencies,  requires  seven  intermediate  relay  stations,  spaced  on  the  average  about 
30  miles  apart  and  located  on  relatively  high  elevations,  as  shown  in  Fig.  1.  This  trial  of 


/P1TTSFIELD 


ASNEBUMSKIT  MT. 
1395' 

/>/ 


II  MILES.  • 

s&  /    "" 


SPRIMGFJELD 

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CONN. 


SQUARE  BLDG. 

(NEWENG-T.&T.CO.) 

ELEV.  *"*-' 


JACKIE 
JONES  MT. 
ELEV.1240y 


FIG.  1.    New  York-Boston  Radio  Relay  System  (Courtesy  Bell  System) 

microwave  transmission  is  for  the  purpose  of  determining  its  efficiency,  dependability, 
and  economy  for  multiplex  telephony  and  for  interconnecting  sound  broadcast  and  tele- 
vision stations,  and  its  application  in  the  network  of  communications  routes.  The  very 
broad  band  of  frequencies  available  for  experimental  use  in  the  super  high-frequency 
(SHF)  range,  coupled  with  the  fact  that  these  waves  tend  to  travel  in  a  straight  line  and 
are  capable  of  being  beamed  by  means  of  guiding  lenses  and  reflectors,  provides  a  promis- 
ing field  for  experiments  in  communications,  with  the  probability  of  eventually  securing 


17-56 


TELEPHONY 


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APPLICATIONS 


17-57 


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From  plane 

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From  train  

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,  
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o£  I 

17-58 


TELEPHONY 


large  groups  of  telephone,  telegraph,  television,  sound  broadcast,  and  other  useful  chan- 
nels. 

The  multicavity  magnetron,  a  high-frequency  power  generator  developed  for  radar, 
has  made  possible  10,000-Mc  frequency  currents  with  peak  powers  ranging  from  10  to 
1000  kw  for  very  short  intervals  of  time. 

The  microwave  radio  relay  system  has  been  given  a  valuable  tool  in  the  lens-antenna, 
consisting  of  an  array  of  small  metal  plates  mounted  in  a  frame  about  10-ft  square.  This 
lens  employs  the  same  principle  in  focusing  radio  waves  into  a  pencil  beam  as  an  optical 

TRANSMITTER   AND 
SWITCHING   CIRCUIT 


CARRJER- 

-=r  OPERATED 
RELAY 

TO  ADDITIONAL  RECEIVER  CIRCUITS 
FIG.  2.     Shore  Circuit  for  Coastal  Radio  Service  (Courtesy  Bell  System) 

lens  does  in  directing  light  waves.  The  radio  waves  are  fed  into  this  new  lens-antenna 
through  a  hollow-tube  wave  guide  and  horn  at  the  rear,  the  waves  spreading  out  along 
the  hornlike  shields  to  the  lens  plates,  which  bend  the  waves  and  direct  them  out  of  the 
front  of  the  lens  to  produce  an  emergent  wave  front  parallel  to  the  front  of  the  lens.  This 
type  of  antenna  causes  less  wave  distortion  due  to  dimensional  variations  than  would  re- 
sult from  such  variations  in  a  parabolic  reflector,  and  greater  manufacturing  tolerances 
therefore  are  permissible.  Also,  the  horn-type  shield  at  the  rear  of  the  lens  reduces  rear- 
ward radiation,  present  in  reflector  antennas,  and  the  front  of  the  lens  may  be  protected 
from  weather  by  a  plastic  covering. 

Multiplex  operation  over  microwave  radio  channels  was  developed  by  the  Bell  Labora- 
tories during  World  War  II  for  use  of  the  TJ.  S.  Army  Signal  Corps  and  proved  to  be  of 
great  value  to  the  armed  forces.  Its  full  possibilities  for  commercial  service  are  being 
explored.  Army-type  pulse  modulated  microwave  systems  are  now  in  commercial  service 
between  Los  Angeles  and  Catalina  Island  as  well  as  from  the  mainland  to  Nantucket  Is- 
land. As  developed  for  war  purposes,  multiplex  operation  employed  a  highly  directive 


APPLICATIONS 


17-59 


and  sharply  focused  microwave  beam  of  about  5000  Me  which  carried  eight  separate  mes- 
sages. The  intelligence  of  each  channel  is  conveyed  by  varying  the  time  position  of  the 
1-microsecond  channel  pulses,  eight  of  which  (one  for  each  channel)  are  transmitted  in 
sequence,  8000  times  a  second.  Thus,  if  a  1000-cycle  tone  is  being  beamed  over  one  chan- 
nel, one  cycle  of  this  tone  requires  1/iooo  of  a  second,  during  which  time  eight  pulses,  spaced 
at  approximately  equal  intervals  throughout  the  one  cycle,  would  be  transmitted.  At 
the  receiving  terminal  these  pulses  are  received  in  sequence  by  their  respective  channel 
and,  being  representative  of  the  electrical  intelligence  at  the  originating  terminal,  are  re- 
converted into  sound  intelligence.  Two- 
way  operation  is  obtained  by  using  a 
separate  radio  channel  in  each  direction 
of  transmission.  Because  of  the  method 
of  transmitting,  this  system  has  been 
designated  as  pulse-position  modulation 
(PPM}. 

Coastal  harbor  and  inland  waterways 
radio  telephone  systems  are  extensive, 
one  consisting  of  14  basic  shore  stations, 
strategically  located  to  cover  the  entire 
coast  from  Maine  to  Florida,  to  Texas, 
and  up  the  Pacific  coast  to  Seattle.  Fig- 
ure 2  shows  a  circuit  of  one  of  the  shore 
stations.  Note  that  by  use  of  the  mon- 
itor jacks  the  shore  operator  can  select 
the  receiver  giving  the  best  reception. 

Highway  mobile  radio  telephone  sys- 
tems give  the  operator  the  same  facility. 
These  systems  consist  principally  of 
(1)  f-m  (frequency  modulation)  radio 
transmitters  (about  50  to  100  miles 
apart)  and  associated  receivers  at  fixed 
locations  along  the  intercity  highway  which  the  system  is  to  cover,  (2)  f-m  radio  trans- 
mitters and  receivers  for  the  mobile  units,  and  (3)  a  control  terminal  associated  with  each 
fixed  transmitter. 

Wire  lines  connect  the  fixed  receivers  to  the  control  tenninal,  which  provides  for  link- 
ing the  transmitter  and  as  many  as  eight  receivers  to  a  two-way,  two-wire  line  to  the 
central  office  handling  the  system. 

In  operation,  a  customer  desiring  connection  with  one  of  his  mobile  units  moving  along 
a  highway  between  his  city  and  a  distant  point  which  is  covered  by  a  radio  system  asks 
for  the  mobile  service  operator  in  his  city.  This  operator  has  access  over  a  wire  circuit 

to  the  various  radio  transmitters  along  the 
highway  and,  on  the  basis  of  the  probable 
location  of  the  desired  mobile  unit  as  fur- 
nished by  the  customer,  a  code  assigned 
to  the  particular  vehicle  is  dialed  and  is 
transmitted  by  wire  and  by  the  selected 
radio  transmitter  to  the  called  vehicle. 
This  code  activates  a  selector  set  in  the 
vehicle,  which  gives  a  visual  and  audible 
alarm.  In  casing  from  the  vehicle,  the 
occupant  removes  his  handset  from  its 
holder  and  presses  his  push-to-talk  hand- 
set button,  which  causes  his  transmitter 
to  send  out  a  signal.  This  signal  is  re- 
ceived by  the  nearest  fixed  receiver  along 
the  route  being  traveled  by  the  vehicle, 
which  converts  the  radio-  to  a  voice-fre- 


FIG.  3. 


Mobile  Radio  Transmitter  and  Receiver  in 
Auto  Trunk  (Courtesy  Bell  System) 


FIG.  4.     Control  Unit  and  Hand  Set  of  a  Mobile 

Radio  Unit  Mounted  under  Auto  Instrument  Panel 

(Courtesy  Bell  System) 


quency  signaL  This  signal  then  passes  along  the  receiving  wire  circuit  to  the  terminal 
equipment  at  or  near  the  telephone  office  and  signals  the  mobile  service  operator. 

Figure  3  shows  a  mobile  installation  of  a  transmitter  and  receiver  in  the  trunk  compart- 
ment of  an  automobile;  Fig.  4,  a  control  unit  and  handset  under  the  dashboard  of  an 
automobile;  and  Fig.  5,  a  schematic  of  a  two-tone  selective  signaling  mobile  unit. 

Urban  mobile  radio  telephone  systems  employ  equipments  like  those  in  the  highway 
mobile  service  for  transmitting  and  receiving  telephone  messages  or  signals.  However, 
since  such  systems  are  intended  to  cover  only  an  individual  city  and  its  adjacent  territory 


17-60 


TELEPHONY 


(up  to  about  20  to  25  miles  from  the  transmitter) ,  usually  only  one  transmitting  station 
is  provided  at  a  central  location  in  the  city  and  a  number  of  receiving  stations  are  strate- 
gically placed  within  the  area  to  be  served,  so  that  the  mobile  transmitter  signals  will  be 
received  and  carried  to  the  central  office  in  the  city,  wherever  the  vehicle  may  be  in  the 
area.  One  additional  feature  not  provided  in  highway  service  is  a  one-way  signaling  serv- 
ice: when  certain  signals  are  received  by  the  occupant  of  the  vehicle,  certain  action  is  to 
be  taken.  Voice  communication  is  not  given  in  signaling  service. 

.  The  fixed  f-m  transmitter  will  generally  have  a  power  output  of  250  watts  using  an  as- 
signed frequency  in  the  152-162  Me  band.  The  mobile  f-m  transmitter  will  usually 
have  a  15-watt  output  at  a  different  frequency  in  this  band.  Initially,  mobile  units  will 
be  limited  to  one  channel  for  a  given  city,  and  a  number  of  such  units  will  be  assigned  to 
the  same  channel. 

Selective  signaling  units,  actuated  by  coded  dial  pulses  transmitted  from  the  fixed  trans- 
mitter and  installed  in  the  vehicles,  will  insure  signaling  only  the  vehicle  being  called.  A 
two-letter,  five-digit  code  will  be  used,  such  as  WU  2-5556,  which  will  not  be  duplicated 


OSCILLATORS 


FIG.  5.     Two-tone  Selective  Signaling  System  for  Mobile  Radio  Service  (Courtesy  Bell  System) 

on  the  same  frequencies,  where  interference  might  result.  Since  the  mobile  selector  set 
operates  the  calling  signal  on  the  twenty-third  pulse,  the  sum  of  the  digits  (not  including 
letters)  will  always  add  up  to  23,  unless  the  plan  of  selection  is  later  changed.  Differ- 
entiation between  vehicles  operating  only  in  one  local  area  and  those  that  may  operate  in 
more  than  one  such  area  will  be  effected  by  assigning  different  code  letters  to  the  two 
classes  of  vehicles. 

Railway  mobile  radio  telephone  systems  are  now  (1949)  under  trial  tests  by  several 
railroad  companies  (1)  between  trains  and  railway  control  points,  (2)  within  the  train 
itself,  and  (3)  for  use  of  the  passengers  in  talking  to  fixed  telephones. 

The  last-mentioned  service  makes  use  of  highway  mobile  installations,  which  in  most 
cases  parallel  the  right-of-way.  Service  between  passengers  on  moving  trains  and  land 
telephone  systems  may  be  rendered  by  carrier-induction  methods,  rather  than  by  radio, 
or  by  both  types  of  transmission,  as  may  seem  best  in  the  future. 

Rural  subscriber  radio  telephone  systems  are  now  in  the  trial  stage,  employing  assigned 
frequencies  within  the  152  to  162  Me  band  and  low-power  transmitters. 

At  Cheyenne  Wells,  Colo.,  a  system  has  been  installed  (operating  on  a  trial  basis  hi 
the  frequency  range  of  44  to  50  Me)  which  provides  for  rural  radio  service  to  eight  ranches 
located  from  11  to  21  miles  from  the  town.  None  of  these  ranches  are  reached  by  either 
power  or  wire  telephone  lines,  the  operating  power  for  the  ranch  radio  sets  being  obtained 
from  home  electric  plants.  Four  of  the  ranches  are  served  by  direct  radio  links  to  the 
Cheyenne  Wells  central  office;  the  other  four  are  reached  by  relatively  short  wire  ex- 
tensions from  a  nearby  radio-equipped  ranch.  At  the  central  office  the  eight  stations 
are  grouped,  through  terminal  equipment,  to  form  an  eight-party  rural  line. 

Radio  equipment  at  each  of  the  first  four  ranches  includes  a  transmitter,  receiver,  a 
telephone  set,  and  two  antennas.  The  10-watt  transmitter  and  associated  receiver  are 
housed  in  a  steel  cabinet,  which  can  be  located  out  of  sight,  with  only  the  telephone  in- 
strument in  view.  The  antennas  are  mounted  on  a  pole  or  atop  one  of  the  ranch  buildings. 

At  Cheyenne  Wells,  the  equipment  includes  a  50-watt  transmitter,  receiver,  and  ter- 
minal equipment  for  associating  the  various  radio  links  with  the  central-office  switchboard. 

Further  studies  will  be  made,  based  on  the  results  of  the  trial  tests,  to  determine  the 
most  practicable  types  of  radio  equipment  to  employ  for  rural  service. 


TRANSMISSION  AND  OPERATIONAL  METHODS        17-61 

Special  emergency  radio  telephone  systems  have  been  in  operation  for  many  years  to 
serve  fire,  police,  forestry,  highway,  utility  companies,  and  a  number  of  other  services. 
Operation  is  of  low  power  output  at  frequencies  generally  within  the  1.6-3.3  and  the 
30-40  Me  bands. 

The  equipment  employed  for  the  mobile  services  is,  in  general,  similar  to  the  types  dis- 
cussed under  coastal  harbor,  highway,  and  urban  mobile  service  with  variations  required 
for  the  particular  type  of  service  involved. 

The  telephone  and  other  wire-using  companies  have  been  using  for  several  years  portable 
emergency  radio  telephone  equipment  to  bridge  gaps  in  open-wire  lines  resulting  from 
sleet,  flood,  or  wind  damage. 

One  type  of  portable  radio  telephone  system  for  emergency  use  employs  a  50-watt 
transmitter  arranged  to  operate  at  one  of  ten  selected  frequencies  within  the  2,0-3.1  Me 
band,  and  a  receiver  adjustable  for  any  frequency  within  the  2.0-3.1  Me  band.  One 
antenna  at  each  terminal  is  switched  between  the  transmitter  and  receiver,  depending  on 
the  direction  of  transmission,  by  a  voice-operated  control  unit.  A  volume  limiter  in  the 
transmitter,  and  automatic  volume  control  and  a  codan  (carrier-operated  device,  anti- 
noise)  in  the  receiver,  regulate  the  transmitter  modulation  and  the  receiver  output  and 
operation  to  provide  uniform  transmission.  Its  operating  range  varies  with  the  type  of 
terrain,  noise,  and  atmospheric  conditions  from  about  25  to  50  miles.  Power  for  the 
terminal  units  is  obtained  from  110-120  volt,  60-cycle  supply.  A  sensitivity  control 
circuit  actuates  the  transmitter  at  a  minimum  level  of  —  47  dbm  (47  db  below  1  milliwatt) 
and  likewise  prevents  line  noise  or  room  noise  from  causing  false  operation  of  the  equip- 
ment. 

9.  TRANSMISSION  AND  OPERATIONAL  METHODS 

Quiescent  transmitter  operation  is  employed  either  to  save  power  or  to  permit  the 
installation  of  a  single  transmitter  and  receiver  at  the  same  location  under  conditions  which 
otherwise  would  prevent  satisfactory  communication.  The  transmitter  is  provided  with 
manual  or  voice-operated  control  means  which  render  it  sufficiently  inactive  during  idle 
intervals  so  that  emission  from  it  does  not  interfere  with  reception.  If  saving  power  is  an 
objective,  the  switching  is  directed  toward  securing  nunimum  power  input  during  qui- 
escent periods. 

Two  sidebands  and  carrier  transmission  is  the  most  commonly  used  method,  owing  to 
the  simplicity  of  signal  generation  and  detection.  It  requires  a  radio  transmission  band 
equal  to  twice  the  highest  audio  frequency  to  be  transmitted.  The  carrier  contains  no 
intelligence-bearing  signal  component  but  simplifies  detection  and  is  useful  for  such  con- 
trol purposes  as  automatic  tuning  and  volume  control  at  the  receiving  end  and  for  the 
operation  of  auxiliary  relays.  The  power  required  to  transmit  the  carrier  is  large  com- 
pared with  that  required  for  the  sidebands,  and  if  the  carrier  is  transmitted  continuously 
it  represents  a  considerable  loss.  In  the  case  of  100  per  cent  modulation  by  a  single- 
frequency  tone,  the  carrier  power  is  twice  the  sum  of  the  power  in  the  two  sidebands. 

Spread  sidebands  and  carrier  transmission  uses  sidebands  displaced  from  their  normal 
positions  in  relation  to  the  carrier  by  an  amount  approximately  equal  to  the  audio  band 
transmitted.  It  is  sometimes  used  when  inverter-type  privacy  is  employed,  because  the 
spreading  then  can  be  accomplished  without  additional  modulating  equipment  and  merely 
requires  that  the  inverter  be  designed  for  different  input  and  output  frequency  bands. 
Advantage:  less  stringent  distortion  requirements  are  imposed  on  radio  equipment  be- 
cause predominant  intermodulation  products  fall  outside  the  used  band.  Disadvantage: 
communication  band,  much  wider  than  otherwise  necessary,  is  occupied  inefficiently. 

Two  sidebands,  suppressed  carrier,  transmission  has  not  been  used,  because  the  diffi- 
culties of  correctly  maintaining  the  phase  of  the  reintroduced  carrier  at  the  receiving  end 
generally  offset  the  advantage  of  saving  hi  power  capacity  at  the  transmitter.  Further- 
more, suppression  of  one  sideband  as  weU  as  the  carrier  usually  offers  additional  advantages 
with  less  stringent  synchronization  requirements,  since  in  this  case  the  frequency  of  the 
reintroduced  carrier  can  depart  from  the  correct  value  by  a  few  cycles  without  appreciable 
mutilation  of  speech  quality,  and  it  can  be  off  as  much  as  20  cycles  without  noticeably  re- 
ducing articulation. 

Single  sideband,  suppressed  carrier,  transmission  has  certain  important  advantages 
which  are  especially  valuable  in  long-wave  systems  where  transmitter  power  capacity, 
communication  band  width,  and  static  interference  are  controlling  factors.  These  ad- 
vantages, referred  to  double  sideband  and  normal  carrier  modulated  100  per  cent,  are: 
6-db  increase  in  intelligence-bearing  signal  for  same  transmitter  amplitude  capacity, 
3-db  reduction  in  random  received  noise,  no  emission  from  transmitter  during  idle  periods, 
and  radio  transmission  band  no  greater  than  audio  band  width.  Disadvantages:  more 


17-62 


TELEPHONY 


complicated  modulating  process;  precision  frequency  control  is  required  in  order  to  re- 
introduce  at  the  receiver  the  carrier  suppressed  at  the  transmitter;  transmitter  power 
supply  is  subjected  to  large  load  variations  at  syllabic  frequencies.  This  method  is  em- 
ployed in  the  long-wave  and  short-wave  systems  between  New  York  and  London  and  on 
some  radiotelephone  systems.  In  the  short-wave  system,  synchronization  of  the  re- 
introduced  carrier  at  the  receiving  end  is  accomplished  satisfactorily  by  transmission  of  the 
carrier  or  an  equivalent  pilot  signal  reduced  10  to  20  db  below  normal  carrier. 

Privacy  is  usually  achieved  by  modifying  the  signals  in  one  or  more  ways  which  render 
the  message  substantially  unintelligible  unless  received  with  special  equipment.  Devices 
called  speech  inverters  have  been  developed  which  reverse  the  sequence  of  the  audio  fre- 
quencies before  modulation  in  the  radio  transmitter;  rein  version  is  then  necessary  after 
detection  at  the  receiver.  Additional  privacy  is  obtained  by  varying  the  radio  carrier  in 
a  cyclic  manner.  A  more  complicated  method,  known  as  split-band  privacy,  involves 
dividing  the  total  audio  band  into  several  narrow  bands.  These  can  be  inverted  and/or 
transposed  in  various  arrangements,  and  the  combination  can  be  changed  frequently. 
All  these  methods  have  been  applied  successfully  to  radio  circuits. 


10.  PRINCIPLES  OF  TWO-WAY  OPERATION 

SYSTEMS  WITH  FOUR- WIRE  TERMINALS.  Since  the  emission  of  radio  waves 
and  their  subsequent  detection  at  a  distant  point  constitute  inherently  a  unilateral  proc- 
ess, duplex  operation  requires  two  one-way  radio  systems  acting  in  opposite  directions. 
Such  an  arrangement  for  telephony  is  shown  in  Fig.  6.  At  each  end  of  the  system  a 


Radio  Transmitte 


Radio  Receiver 


Radio  Receiver-^ 


dio  Transmitter 


FIG.  6.     Simple  Radio  System — Four-wire  Terminals 

microphone  is  connected  to  a  radio  transmitter  and  a  telephone  receiver  is  connected  to  a 
radio  receiver.  Thus,  two  persons  can  converse  provided  the  radio  transmission  east- 
ward does  not  interfere  with  the  transmission  westward;  i.e.,  radiation  from  T2  reaching 
RI  must  not  prevent  RI  from  functioning  satisfactorily  in  receiving  signals  from  TI.  Like- 
wise radiation  from  TI  at  RZ  must  not  interfere  with  reception  of  signals  from  T%.  For 
the  condition  where  the  person  at  W  is  talking  and  the  person  at  E  is  listening,  the  effect 
of  Tz  at  RI  may  be  (a)  increased  noise,  (6)  distortion  of  signals  from  TI,  (c)  detection  of 
unwanted  signals.  Furthermore,  the  effect  of  TI  at  #2  may  result  in  a  return  signal  to 
W  which  is  quite  disconcerting.  If  the  person  at  W  hears  himself  talking  in  reasonable 
volume,  it  will  not  disturb  him  unless  there  is  sufficient  delay  to  give  an  echo  effect.  How- 
ever, if  he  hears  unintelligible  sounds  in  like  volume,  they  will  seriously  disturb  him. 
Such  sounds  appear  if  RZ  is  overloaded  by  high  field  intensities  from  TI  or  if  TI  produces 
extra-band  radiation  within  the  selectivity  band  of  RZ  and  of  sufficient  intensity  to  be 
detected. 

The  forms  of  interference-  outlined  above  are  prevented  by  adapting  one  or  a  combina- 
tion of  the  following  expedients:  (a)  use  of  different  frequency  bands  for  the  two  direc- 
tions of  transmission,  (6)  geographical  separation  of  the  radio  transmitter  and  receiver, 
(c)  use  of  directional  antennas,  (d)  suppressing  all  emission  from  transmitters  except  during 
transmission  of  wanted  signal  and  disabling  and  protecting  local  receivers  during  active 
intervals  of  adjacent  transmitter.  Accomplishing  (d)  involves  switching  operations. 
Manual  control  of  the  switching  is  satisfactory  for  experienced  talkers,  but  voice-current 
control  is  favored  for  more  general  use.  Single  sideband,  suppressed  carrier  systems  are 
less  susceptible  to  interference  than  other  systems  and  may  be  used  more  readily  without 
switching  arrangements,  if  the  transmitter  and  receiver  are  separated  sufficiently. 


PRINCIPLES  OF  TWO-WAY  OPERATION  17-63 

In  planning  a  specific  system,  there  is  a  wide  variety  of  circumstances,  including  eco- 
nomic factors  and  the  characteristics  of  the  particular  apparatus  to  be  used,  which  de- 
termine the  selection  of  the  method  or  combination  adopted.  There  are  no  clearly  de- 
fined dividing  lines,  but  it  may  be  stated  rather  generally  that  the  use  of  (a}  or  (d)  alone  is 
satisfactory  for  short-distance  working.  All  long-distance  systems  employ  at  least  (a) 
and  (6)  or  (6)  and  (d).  Geographical  separations  range  from  a  few  hundred  feet  to  sev- 
eral hundred  miles.  _  Use  of  (c)  reduces  the  distance  necessary.  With  (a)  the  necessary 
geographical  separation  is  determined  by  the  attenuation  required  to  bring  the  ratio  of 
field  intensities  of  the  wanted  and  unwanted  signals  within  the  limits  of  the  receiver  se- 
lectivity and  overload  characteristics.  With  (<2),  geographical  separation  is  necessary 
for  high-power  transmitters,  to  avoid  exposing  the  receiving  system  to  noise  due  to  spu- 
rious emanations  from  parts  of  the  transmitting  apparatus  or  the  associated  power-supply 
system. 

SYSTEMS  INTERCONNECTING  WITH  TWO-WIRE  EXTENSIONS.  Since  stand- 
ard telephone  subscriber  loops  are  two-wire  circuits  in  which  messages  in  opposite  direc- 
tions traverse  the  same  wire  path,  the  two  oppositely  directed  radio  paths  in  Fig.  6  must 
be  arranged  to  terminate  two-wire.  The  ordinary  hybrid  coil  arrangement  common  in 
telephone  repeaters  and  four-wire  cable  circuits  fails  to  solve  this  problem  except  where 
the  radio  circuit  meets  all  the  requirements  imposed  by  wire  practice  on  corresponding 
wire  circuits.  This  is  seldom  possible  or  economical  on  account  of  difficulties  peculiar  to 
the  radio  paths.  In  wire  systems,  transmission  levels  remain  fixed  within  closely  es- 
tablished limits,  and  signal  volumes  vary  over  a  considerable  range.  In  radio  circuits, 
comparatively  large  variations  in  attenuation  sometimes  occur  in  relatively  short  intervals 
of  time,  except  over  extremely  short-distance  paths,  and  these  tend  to  cause  retransmission 
of  received  signals  at  such  amplitudes  that  severe  echoes  and  even  singing  around  the  two 
ends  of  the  circuit  will  occur  unless  means  are  provided  to  prevent  it.  Furthermore,  for 
all  long-distance  working,  it  is  uneconomical  to  provide  transmitter  capacity  which  will 
permit  appreciable  variations  in  signal  volume.  To  obtain  maTin-mm  signal-noise  ratios 
at  the  radio  receivers,  it  is  essential  that  the  speech  currents  fully  load  the  transmitters. 
This  requires  gain  adjustments  between  the  hybrid  coils  and  the  transmitters  to  suit  the 
particular  talkers  and  the  condition  in  the  connecting  wire  circuits. 

To  overcome  these  fundamental  transmission  difficulties,  automatic  switching  systems 
operated  by  the  voice  currents  of  the  speakers  have  been  developed.  These  devices  block 
the  radio  path  in  one  direction  while  speech  is  traveling  in  the  reverse  direction  and  also 
keep  one  direction  blocked  when  no  speech  is  being  transmitted.  The  operation  is  so 
rapid  that  it  is  unnoticed  by  the  telephone  users.  Since  these  systems  prevent  the  exist- 
ence of  singing  and  echo  paths,  then-  use  permits  the  amplification  to  be  varied  at  several 
points  almost  without  regard  to  changes  in  other  parts  of  the  system,  and  it  is  possible 
by  manual  or  automatic  adjustment  to  maintain  the  volumes  passing  into  the  radio  link 
at  relatively  constant  values  irrespective  of  the  lengths  of  the  connected  wire  circuits  and 
the  talking  habits  of  the  subscribers. 

Figure  7  is  a  schematic  diagram  of  one  end  of  a  circuit  showing  the  essential  features  of 
a  voice-operated  device.  This  kind  of  apparatus  is  capable  of  taking  many  forms  and  is, 
of  course,  subject  to  change  as  improvements  are  developed.  The  diagram  illustrates  how 
one  of  these  forms  might  be  set  up.  This  form  employs  electromechanical  relays.  The 
functioning  of  the  apparatus  illustrated  is  briefly  as  follows:  the  relay  TES  contact  is 
normally  open  so  that  received  signals  pass  through  to  the  subscriber.  The  relay  SS  con- 
tact is  normally  closed  to  short-circuit  the  transmitting  line.  When  the  subscriber  at 
W  speaks,  his  voice  currents  go  into  both  the  transmitting  detector  and  the  transmitting 
delay  circuits.  The  transmitting  detector  is  a  device  that  amplifies  and  rectifies  the  voice 
currents  to  produce  currents  suitable  for  operating  the  relays  TES  and  SS,  which  there- 
upon short-circuit  the  receiving  line  and  clear  the  short  circuit  from  the  transmitting  line, 
respectively.  The  delay  circuit  is  an  artificial  line  through  which  the  voice  currents  re- 
quire a  few  hundredths  of  a  second  to  pass  so  that  when  they  emerge  the  path  ahead  of 
them  has  been  cleared  by  the  relay  SS.  When  the  subscriber  at  W  has  ceased  speaking, 
the  relays  drop  back  to  normal.  The  function  of  the  receiving  delay  circuit,  the  receiving 
detector,  and  the  relay  RES  is  to  protect  the  transmitting  detector  and  relays  against  op- 
eration by  echoes  of  received  speech  currents.  Such  echoes  arise  at  irregularities  in  the 
two-wire  portion  of  the  connection  and  are  reflected  back  to  the  input  of  the  transmitting 
detector,  where  they  are  blocked  by  the  relay  RES  which  has  closed  and  which  hangs  on 
for  a  brief  interval  to  allow  for  echoes  that  may  be  considerably  delayed.  The  gain  con- 
trol potentiometers,  shown  just  preceding  the  transmitting  and  receiving  amplifiers,  are 
provided  for  the  purpose  of  adjusting  the  amplification  applied  to  outgoing  and  incoming 
signals. 

The  relief  from  severe  requirements  on  stability  of  radio  transmission  and  from  varying 


17-64 


TELEPHONY 


speech  load  on  the  radio  transmitters,  which  such  a  system  provides,  permits  much  greater 
freedom,  in  the  design  of  the  two  radio  circuits  than  would  otherwise  be  possible.  In  the 
system  shown  in  Fig.  7' interference  between  local  transmitter  and  receiver,  as  outlined 
previously  in  discussing  Fig.  6,  is  prevented  by  such  geographical  separation  as  may  be 
necessary  in  combination  with  either  the  use  of  two  communication  bands  or  single  side- 
band, suppressed  carrier,  transmission.  When  one  communication  band  is  used  for  both 
directions  with  carrier  and  double  sideband  transmission,  the  switching  systems  of  the 
type  shown  in  Fig.  7  are  extended  to  operate  additional  devices  which  suppress  the  car- 
rier at  the  transmitter  and  disable  the  receiver.  This  switching  is  necessary  to  protect 

Vprom  Distant 
I   Transmitter 


Subscriber     Tefephone 
Operator 


FIG.  7.     Two- wire  Radio  Terminal  Showing  Arrangement  of  Voice-operated  Switching  Devices 

the  receiver  where  it  is  exposed  to  intense  fields  from  the  transmitter.  When  privacy  ap- 
paratus is  included  in  an  installation,  the  voice-frequency  switching  is  frequently  arranged 
to  transfer  the  same  privacy  unit  from  the  transmitting  to  the  receiving  leg  of  the  four- 
wire  terminal  (or  vice  versa) ,  thus  saving  cost  of  a  second  privacy  unit. 


11.  SYSTEM  DESIGN 

The  designer  of  a  radio  circuit  is  limited  to  the  establishment  of  certain  facilities  at  the 
transmitting  and  receiving  station  which  are  expected  to  yield  the  desired  results  on  the 
basis  of  available  data,  computations,  and  previous  similar  experience.  Beyond  this 
point,  the  performance  is  inherently  a  matter  of  probability,  since  transmission  is  subject 
to  the  vagaries  of  natural  influences  entirely  beyond  the  control  of  man.  Sometimes, 
very  short-distance  circuits  can  be  provided  which  are  substantially  immune  from  these 
influences.  However,  the  designers  of  long-distance  circuits  frequently  encounter  tech- 
nical limitations  which  determine  the  maximum  degree  of  reliability  attainable,  quite 
apart  from  considerations  of  cost. 

The  design  proceeds  from  a  statement,  which  includes  (a)  type  and  nature  of  service 
expected,  (6)  daily  hours  of  operation,  (c)  general  location  of  terminals,  (d)  distance  cov- 
ered and  character  of  intervening  region,  (e)  overall  transmission  requirements,  (/)  future 
plans.  From  these  now  the  requirements  and  compromises  which  ultimately  determine 
the  selection  of  transmission  frequencies  and  methods;  the  location  and  general  arrange- 
ment of  transmitting,  receiving,  and  voice  terminal  stations;  and  the  choice  of  equip- 
ment. 

Systems  requiring  separate  transmitting  and  receiving  stations  usually  involve  expend- 
itures which  warrant  a  fairly  comprehensive  preliminary  survey.  The  survey  includes  a 
search  for  suitable  station  sites  and  measurements  of  received  field  intensities,  noise  levels, 
and  angles  of  wave  arrival  over  the  approximate  path  at  substantially  the  proposed  fre- 
quencies. This  survey  frequently  can  be  effected  by  observing  the  signals  from  existing 
telephone  or  telegraph  stations.  It  serves  to  substantiate  conclusions  derived  from 
computations  and  from  available  transmission  data;  it  gives  important  specific  information 
concerning  noise  at  receiving  sites;  and  it  should  serve  to  disclose  any  conditions  peculiar 
to  particular  locations  which  may  have  a  profound  effect  upon  performance. 


SYSTEM  DESIGN  17-65 

TRANSMISSION  REQUIREMENTS  are  related  intimately  with  the  needs  and  objec- 
tives of  individual  applications  to  such  an  extent  that  the  statements  herein  should  not 
be  taken  as  concrete  recommendations  but  should  be  regarded  principally  as  guides  to  the 
items  that  need  consideration  when  formulating  the  specific  requirements  for  a  particular 
system. 

Circuits  terminating  four-wire  and  used  only  for  direct  conversations  between  terminals 
can  be  operated  over  a  wide  scale  of  conditions  all  of  which  may  be  acceptable  for  the 
purpose  in  hand.  Circuits  interconnecting  with  telephone  plants  should  conform  as  far 
as  possible  to  the  standards  of  the  wire  systems  in  respect  to  transmission,  stability,  dis- 
tortion, and  interference  effects.  Allowance  must  be  made  for  similar  imperfect  con- 
ditions in  wire  transmission.  Otherwise  the  complete  connection  may  be  unsatisfactory, 
even  though  it  is  possible  to  converse  over  the  radio  circuit  alone.  With  the  possible 
exception  of  very  short-distance  radio  circuits,  it  is  generally  true  that  radio  transmission 
conditions  vary  through  an  extremely  wide  range,  and  it  becomes  necessary  during  some 
operating  intervals  to  endure  a  comparatively  poor  circuit  or  temporarily  do  without  one. 
It  is  desirable,  therefore,  to  make  the  limiting  requirements  as  liberal  as  possible. 

Transmission  Times.  The  requirements  are  no  different  from  those  for  corresponding 
wire  systems.  The  transmission  time,  from  radio  transmitter  input  terminals  to  receiver 
output  terminals,  is  seldom  appreciably  greater  than  the  time  required  for  radio  waves 
to  traverse  from  transmitter  to  receiver.  The  propagation  rate  is  approximately  186,000 
miles  per  second.  To  this  must  be  added  time  for  wire  circuits  to  the  control  centers, 
time  for  delay  networks  used  with  voice-operated  switching  devices,  and  time  for  any 
delays  incurred  in  other  apparatus,  such  as  privacy  systems. 

Stability.  It  is  desirable  that  all  equipment  adjustments  that  affect  the  overall  circuit 
performance,  and  more  particularly  those  that  cannot  be  made  without  removing  the  cir- 
cuit from  service,  retain  the  established  conditions  for  long  periods. 

Receiving-set  selectivity,  in  addition  to  discriminating  sharply  against  unwanted  sig- 
nals, should  provide  ample  margins  for  the  total  effect  of  frequency  drifts  to  be  tolerated 
over  a  period  of  several  hours  at  all  points  in  the  system  where  base  frequencies  are  gen- 
erated. For  example:  a  system  employing  a  quartz-plate  oscillator  at  the  transmitter 
and  a  well-designed  beating  oscillator  of  the  tuned  circuit  type  at  the  receiver,  when 
operating  at  15  Me  and  transmitting  audio  frequencies  up  to  3000  cycles,  requires  an 
intermediate-frequency  band  width  of  the  order  of  8000  cycles  (— 3  db  at  4000  cycles  from 
midband) . 

When  the  system  is  operated  on  the  basis  of  constant  net  loss,  it  is  desirable  that  the 
loss  in  the  radio  circuits  be  held  to  ±1  db.  When  the  system  is  operated  on  the  basis  of 
constant  input  volume  to  the  radio  transmitter  and  voice-operated  devices  are  used,  vari- 
ations of  d=5  db  are  sometimes  tolerated  at  the  receiving  end.  When  radio  transmission 
is  subject  to  rapid  variations  the  receivers  are  provided  with  automatic  volume  controls 
which  hold  the  output  volume  within  a  few  decibels  for  wide  fluctuations  in  the  signal- 
field  intensity. 

Audio  distortion  is  usually  specified  in  terms  of  transmission  band  characteristics, 
amplitude  or  load  characteristics,  and  departure  from  proportionality  of  phase  shift  with 
frequency.  Difficulties  from  the  last  are  seldom  encountered  in  ordinary  radio  circuits 
used  for  voice  communication. 

Transmission  Band  Characteristics.  The  requirements  in  each  instance  axe  closely 
related  to  the  needs  of  the  particular  application.  Except  when  using  radio  frequencies 
less  than  100  kc,  it  is  seldom  a  difficult  matter  to  provide  reasonable  bands  for  conversa- 
tional purposes.  The  minimum  requirements  for  international  radio  telephone  circuits, 
as  recommended  by  the  International  Consultative  Committee  on  Telephony,  are  the 
same  as  for  long-distance  wire  circuits:  i.e.,  300-2600  cycles  as  the  limiting  frequencies 
effectively  transmitted.  The  committee  also  recommends  that  future  circuits  be  designed 
to  transmit  at  least  200—3000  cycles.  Widening  the  band  improves  quality  and  to  a  lesser 
extent  improves  articulation,  provided  that  other  conditions  do  not  become  controlling. 
For  instance,  exposure  to  random  noise  at  the  radio  receivers  is  substantially  proportional 
to  the  band  width,  so  that  widening  the  band  may  not  result  in  improved  performance 
under  adverse  receiving  conditions.  In  the  United  States,  it  is  customary  to  specify  the 
transmission-frequency  characteristic  in  terms  of  departure  in  decibels  from  the  trans- 
mission level  at  1000  cycles. 

Amplitude  or  Load  Characteristics.  Except  when  radio  circuits  are  subject  to  rapidly 
varying  transmission  conditions,  such  as  those  frequently  encountered  in  the  short-wave 
range,  load  tests  are  made  in  the  same  manner  as  for  wire  circuits,  and  the  requirements 
are  the  same  for  equivalent  results. 

The  linearity  requirements  for  the  transmitting  system  and  the  receiving  system  are 
usually  specified  in  terms  of  distortion  products  resulting  from  the  application  of  pure 


17-66  TELEPHONY 

tones.  The  tests  are  made  separately  for  the  two  systems.  Carrier  and  double  sideband 
systems  may  be  tested  with  a  single  tone  at  varying  amplitudes.  Single  sideband  sys- 
tems require  two-tone  tests.  If  privacy  apparatus  is  to  be  used  with  double  sideband 
systems,  it  is  usually  required  that  the  total  distortion  products  introduced  by  the  trans- 
mitter alone  or  the  receiver  alone  and  falling  within  the  audio  band  shall  be  about  25  or 
30  db  below  the  single  tone  output  for  any  tone  input  up  to  90  per  cent  modulation.  If 
two  equal  tones  are  used  on  any  system,  it  is  generally  specified  that  any  distortion  product 
falling  within  the  transmission  band  shall  be  20  to  25  db  below  one  tone.  (For  testing 
methods,  see  Section  11  and  I.R.E.  Report  of  Standardization  Committee.) 

Radio-frequency  or  phase  modulation  at  the  transmitters  results  in  troublesome  distor- 
tion effects  at  the  receiving  station  if  radio  transmission  occurs  simultaneously  over  two 
or  more  paths  differing  in  length  by  appreciable  portions  of  the  wavelength  used.  This 
situation  is  encountered  in  the  short-wave  systems,  and  it  is  sometimes  specified  that, 
during  the  modulation  cycle,  the  phase  shift  associated  with  frequency  modulation  or 
phase  modulation  should  not  exceed  ±15°.  (For  method  of  test,  see  I.R.E.  Report  of 
Standardization  Committee.) 

When  no  appreciable  differences  in  simultaneous  transmission  path  lengths  are  en- 
countered, frequency  modulation  can  be  tolerated  if  all  the  radio  equipment  has  sub- 
stantially a  flat  frequency-transmission  characteristic  throughout  a  band  sufficiently  wide 
to  pass  all  the  essential  frequencies  generated  and  if  extra-band  radiation  does  not  inter- 
fere with  other  services. 

Interference  is  caused  by  signals  from  other  radio  circuits  and  by  disturbances  generally 
classified  as  noise. 

Unwanted  signals  may  enter  the  radio  circuit  through  cross-modulation  effects  at  the 
transmitting  station  if  there  are  two  or  more  transmitters,  or  they  may  enter  at  the  re- 
ceiver. Cross-modulation  is  likely  to  occur  with  open-wire  transmission  lines.  It  is  not 
difficult  to  overcome  if  active  lines  are  well  separated  and  long  parallel  runs  are  avoided. 
It  results  from  the  impression  on  the  tube  circuits  of  one  transmitter  of  modulated  radio- 
frequency  voltages  generated  by  a  second  transmitter  coupled  to  the  first  usually  via  the 
transmission  lines  or  the  antennas.  In  special  cases,  trap  circuits  or  other  simple  filtering 
devices  are  introduced  when  found  necessary,  but  they  are  objectionable  if  the  same  lines 
are  to  be  used  for  several  frequency  assignments. 

At  the  receiving  station  it  is  not  enough  to  provide  apparatus  having  comparatively  high 
selective  properties.  It  is  necessary  to  know  in  what  manner  this  selectivity  is  achieved, 
and  what  values  of  unwanted  signal  voltage  may  be  impressed  upon  the  receiver  input- 
terminals  simultaneously  with  the  wanted  signal.  A  receiving  site  and  an  antenna  sys- 
tem must  then  be  selected  which  will  not  violate  these  receiver  requirements. 

Noise  at  the  receiving  terminal  is  derived  from  the  connecting  wire  system,  the  radio 
transmitting  and  receiving  apparatus,  and  the  radio  noise  field.  If  noise  from  the  wire 
system  meets  the  accepted  standards  for  good  toll  circuits,  as  it  should,  it  is  not  likely  to 
have  a  noticeable  effect  on  the  performance  of  the*  radio  circuit.  Noise  generated  within 
the  radio  transmitter  is  measured  in  terms  of  audio  signal  by  means  of  a  linear  monitoring 
rectifier  exposed  to  the  transmitter  radio  output.  The  audio  signal-noise  ratio  thus  ob- 
tained should  be  somewhat  better  than  the  maximum  audio  signal-noise  ratio  which  it  is 
desired  to  obtain  at  the  receiving  end  under  conditions  of  high  signal-field  and  low  noise- 
field  intensities.  Noise  due  to  the  receiving  equipment  should  never  be  a  controlling 
factor  except  when  approaching  the  limit  of  sensitivity. 

Since  the  effect  of  noise  depends  greatly  on  its  frequencies  in  relation  to  the  audio  trans- 
mission band,  precautions  are  necessary  in  systems  employing  frequency  inversions  to 
prevent  the  conversion  of  relatively  harmless  noise  into  very  objectionable  noise.  This 
conversion  may  occur  if  the  noise  enters  any  part  of  the  system  between  points  where  the 
inversions  and  reinversions  are  made. 

The  interfering  effect  of  noise  is  very  difficult  to  express  accurately. 

12.  INSTALLATIONS 

The  successful  establishment  and  maintenance  of  dependable,  long-distance  circuits 
with  two-wire  terminations  require  careful  installation  planning,  the  provision  of  adequate 
testing  facilities,  and  the  consideration  of  many  problems  only  indirectly  related  to  the 
technical  operation  of  the  system.  There  is  a  wide  gap  between  this  extreme  and  the 
simple  facilities  required  for  short-distance  radio  circuits  without  two-wire  extensions 
(Section  7). 

TRANSMITTERS  AND  RECEIVERS  AT  SAME  LOCATION.  Small  transmitting 
systems  for  short-distance  service  are  placed  at  the  same  location  as  the  receiving  sys- 


INSTALLATIONS  17-67 

terns.  The  transmitter  and  receiver  are  usually  self-contained  units  requiring  a  single 
connection  to  the  general  power  supply.  If  the  same  frequency  assignment  is  used  for 
both  directions  of  transmission  a  single  antenna  is  sufficient.  Manual  or  voice-controlled 
switching  is  necessary  to  change  from  receive  to  transmit  conditions.  If  two  frequency 
assignments  are  used  without  quiescent  transmitter  operation,  it  is  frequently  found  more 
satisfactory  to  employ  two  antennas  slightly  separated  than  to  make  provision  for  trans- 
mission and  reception  on  the  same  antenna.  The  latter  is  possible  by  means  of  various 
special  circuit  arrangements  but  is  almost  certain  to  incur  a  penalty  in  respect  to  min- 
imum receivable  field  intensities.  Since  the  field  intensity  gradient  around  the  trans- 
mitting antenna  is  extremely  steep,  it  is  seldom  necessary  to  remove  the  receiving  antenna 
more  than  50  to  500  ft  in  order  to  secure  satisfactory  conditions.  The  distance  depends  on 
transmitter  power,  type  of  antenna,  frequency  difference,  receiver  selectivity,  and  load 
characteristics.  If  the  antennas  have  directional  properties,  the  relative  positions  should 
be  selected,  when  possible,  so  that  each  antenna  presents  a  null  in  the  direction  of  the 
other.  Usually  the  transmitting  antenna  is  erected  near  the  apparatus,  and  the  receiving 
antenna  is  placed  at  a  distance  from  the  receiver,  connections  being  provided  by  suitable 
transmission  lines. 

If  it  is  essential  that  the  transmitting  and  receiving  apparatus  be  installed  in  close 
proximity,  attention  needs  to  be  given  to  shielding  to  prevent  direct  interference  between 
transmitter  and  receiver.  Receivers  designed  for  this  type  of  installation  seldom  require 
further  shielding  when  used  with  transmitters  up  to  about  25-watt  capacity,  provided  that 
the  transmitters  are  also  reasonably  well  shielded.  It  is  a  good  plan  to  place  receivers 
somewhat  away  from  transmitters  for  ratings  up  to  about  500  watts.  The  alternative  is 
to  provide  a  special  shielded  compartment  for  the  receiver.  This  has  been  done  on  ship- 
board, where  space  limitation  and  operating  convenience  demand  a  compact  installation. 

As  the  transmitter  power  is  increased,  the  possibilities  increase  rapidly  that  noise  will 
enter  the  receiver  directly  or  through  the  receiving  antenna  from  various  sources  within 
the  transmitter  or  its  power  circuits.  If  this  occurs,  it  limits  the  permissible  receiver 
sensitivity  and  may  completely  nullify  the  value  of  higher  power  for  the  purpose  of  work- 
ing greater  distances.  Recourse  to  voice-controlled  quiescent  transmitter  operation 
greatly  alleviates  this  type  of  interference  for  installations  where  two  or  more  transmitters 
are  not  to  be  used  simultaneously.  It  is  frequently  applied  to  ship  systems  and  ma- 
terially increases  the  working  distances.  It  is  effective  only  hi  eliminating  noises  related 
to  the  suppressed  radio  signal  components.  Around  large  transmitters  the  residual  noise 
after  the  carrier  or  other  radio  signal  components  are  suppressed  still  prevents  the  use  of 
extremely  sensitive  receivers.  This  is  one  of  the  compelling  reasons  for  establishing 
separate  transmitting  and  receiving  stations  for  long-distance  circuits,  where  extremes  of 
power  and  sensitivity  are  essential. 

SELECTION  OF  TRANSMITTING  STATION  SITE.  Items  requiring  consideration 
are:  ground  conductivity  and  dielectric  constant;  general  character  of  surrounding  ter- 
rain; position  relative  to  receiving  stations;  possibilities  of  interfering  with  broadcast  re- 
ception or  that  of  other  services;  transportation,  power,  and  telephone  facilities;  living 
arrangements  for  station  personnel;  prevalence  of  sleet  storms;  unusual  conditions  of  tem- 
perature, humidity,  presence  of  salt  spray;  etc.  Ground  conditions  affect  antenna  design. 
Desirable  characteristics  depend  upon  the  type  of  radiating  system  to  be  employed,  the 
frequencies  and  the  wave  angles  of  transmission  (Sections  6  and  10).  The  terrain  in  the 
direction  of  transmission  affects  the  vertical  wave  angle.  Mountains  and  bills  subtending 
large  angles  are  undesirable.  Steel  towers,  buildings,  transmission  lines,  etc.,  constituting 
sizable  obstructions  directly  in  front  of  short-wave  directional  antennas,  are  objectionable 
since  they  modify  the  directive  pattern. 

THE  TRANSMITTING  STATION  LAYOUT  is  based  primarily  on  the  requirements  of 
the  antennas  and  their  relation  to  the  transmitter.  Usually  the  antenna  or  antennas  are 
located  a  short  distance  away  from  the  building  housing  the  transmitter,  and  connection 
is  made  through  open-wire  or  concentric  transmission  lines.  The  practice  of  bringing 
the  antenna  downlead  directly  to  the  transmitter  is  seldom  followed  in  modern  installa- 
tions. Use  of  uniform  transmission  lines,  all  having  the  same  impedance,  greatly  simplifies 
switching  problems.  Good  values  are:  open-wire  600  ohms,  concentric  lines  70  to  80 
ohms.  Placing  the  antennas  clear  of  the  transmitter  building  avoids  difficulties  in  erec- 
tion and  maintenance.  Short-wave  directional  antennas  are  placed  so  that  the  building 
is  not  within  the  horizontal  angle  of  the  principal  lobe. 

RECEIVING  STATION  EQUIPMENT  depends  somewhat  on  the  number  of  radio 
circuits  involved  but  is  also  influenced  considerably  by  the  standards  established  in  re- 
spect to  service  interruptions  other  than  those  attributable  to  the  transmitting  medium. 
The  essential  components  are:  radio  receiver  and  its  power-supply  units,  antennas  and 
transmission  lines,  wire-terminal  apparatus  and  voice-frequency  testing  equipment  with 


17-68  TELEPHONY 

associated  power-supply  units,  general  power  supply,  including  emergency  power  sources. 
Large  stations  usually  have  facilities  for  observing  the  field  intensities  of  received  signals 
and  noise  and  for  the  precise  measurement  of  received  frequencies.  These  are  used  to 
cheek  the  transmitters  which  are  a  part  of  the  system,  and  those  of  other  systems,  which 
create  interference. 

SELECTION  OF  RECEIVING  STATION  SITE  is  a  matter  demanding  careful  con- 
sideration, especially  if  long-distance  services  are  contemplated,  and  it  is  seldom  safe  to 
make  a  final  decision  without  actual  observations  of  signal-field  and  noise-field  intensities 
over  a  period  sufficient  to  obtain  representative  data.  Items  that  should  receive  attention 
are:  location  relative  to  transmitting  stations  of  same  system  and  all  other  nearby  trans- 
mitters and  sources  of  man-made  noise;  local  ground  conditions  and  general  character  of 
surrounding  terrain  in  the  direction  of  wave  arrival;  transportation,  power,  and  telephone 
facilities;  living  arrangements  for  station  personnel;  prevalence  of  electrical  storms;  un- 
usual conditions  of  temperature,  humidity,  presence  of  salt  spray,  etc. 

It  is  desirable  to  have  the  antennas  present  nulls  toward  all  transmitters  in  the  area 
which  are  likely  to  produce  any  form  of  interference. 

Likely  sources  of  man-made  interference  are:  high-tension  transmission  lines,  electrical 
machinery  in  factories,  electrical  trains,  automobiles,  airplanes,  motorboats,  etc.  The 
last  three  mentioned  are  particularly  important  in  long-distance  short-wave  reception  at 
times  when  signal-field  intensities  are  low  as  the  result  of  magnetic  disturbances.  With 
an  extremely  sensitive  receiver  and  a  directional  antenna  designed  for  low-angle  reception, 
no  serious  interference  would  be  expected  from  automobiles  1  1/2  miles  in  front  of  antenna 
and  1/2  to  !/4  mile  at  the  sides  and  rear. 

Reception  of  short  waves  arriving  at  low  angles  can  frequently  be  improved  from  5  to 
10  db  by  placing  properly  designed  antennas  on  ground  sloping  uniformly  downward  in 
the  direction  of  the  transmitting  station  at  an  angle  of  5°  to  15°. 

RECEIVING  STATION  LAYOUT.  Primary  objectives  are  to  place  the  antennas  in 
an  advantageous  position  for  the  collection  of  energy  from  the  incoming  waves  and  to 
obtain  an  efficient  arrangement  in  respect  to  transmission  lines.  In  choosing  directional 
antenna  locations,  close  attention  should  be  given  to  the  position  of  objects  capable  of 
reflecting  or  otherwise  redirecting  unwanted  waves  into  the  sensitive  angles  of  the  antenna 
characteristic. 

It  is  well  to  adopt  a  uniform  impedance  for  all  transmission  lines.  Convenient  values 
are:  open  wire  600  ohms,  concentric  lines  70  to  80  ohms.  In  order  to  avoid  disturbing 
the  incoming  radio  waves,  and  also  to  avoid  undesirable  currents,  transmission  lines 
should  be  placed  as  near  the  ground  as  practicable.  However,  it  is  inadvisable  to  place 
two-wire  lines  less  than  6  ft  from  the  ground.  Pour-wire  balanced  lines  are  disturbed  less 
by  the  proximity  of  the  ground  and  have  been  used  successfully  at  4-ft  elevations.  Con- 
centric lines  may  be  installed  underground.  They  should  always  be  sealed  and,  in  some 
situations,  are  further  protected  from  moisture  by  maintenance  of  pressure  with  inert  gas. 
Aside  from  somewhat  higher  first  cost,  much  can  be  said  in  favor  of  concentric  conductors, 
since  they  substantially  eliminate  the  difficulties  encountered  with  converging  lines  at  re- 
ceiving set  locations.  At  short-wave  stations,  it  is  desirable  to  have  all  directional  an- 
tennas present  a  null  to  the  building  and  the  road  approaching  it. 

SHIP  STATIONS  are  usually  installed  in  extremely  limited  quarters.  They  require 
compact  units,  designed  to  allow  inspection  and  repairs  without  having  access  to  all  sides 
and  preferably  without  disconnection  and  removal  of  parts.  Rapid  frequency-changing 
features  are  especially  important  if  it  is  desired  to  maintain  close  contact  with  more  than, 
one  shore  station, 

Transmittuig  and  receiving  antennas  are  usually  separated  as  much  as  possible.  Simple 
types  are  generally  used  because  they  are  suitable  for  several  frequencies.  Horizontal 
directional  properties  are  not  desirable.  Electrical  noise  conditions  vary  widely  with 
positions  aboard  ship. 

A  four-wire  termination  is  usually  employed  to  avoid  rather  expensive  control-office 
equipment.  Ship's  passengers  talk  from  a  conveniently  located  booth.  Circuits  are  also 
provided  from  the  captain's  quarters  or  a  similar  point  convenient  to  the  bridge.  With  a 
four-wire  terminal,  no  voice-operated  switching  apparatus  is  needed  other  than  the  simple 
devices  necessary  for  quiescent  transmitter  operation.  Voice-frequency  apparatus  is 
mounted  adjacent  to  the  receiver.  One  attendant  supervises  all  operations  and  performs 
the  duties  of  a  technical  operator.  If  the  ship  is  equipped  with  radio  telegraph  as  well 
as  telephone  apparatus,  and  the  two  must  operate  simultaneously,  precautions  are  neces- 
sary to  avoid  mutual  interference. 

SHORE  STATIONS  are  usually  equipped  to  offset,  as  far  as  practicable,  unfavorable 
conditions  aboard  ship.  This  is  done  by  providing  transmitter  power  capacity  from  10 
to  40  times  that  of  the  ship  station,  by  employing  directional  antennas,  and  by  selecting 


SERVICE  REQUIREMENTS — TOLL  17-69 

a  quiet  receiving  location  a  few  miles  away  from  the  transmitting  station.  Antennas 
having  moderately  directional  patterns  (6  to  12  db  maximum  net  gain)  are  used  to  cover 
the  principal  ship  lanes  effectively.  Less  directional  antennas  are  needed  for  general 
coverage.  When  a  ship  is  close  in,  at  "which  time  the  required  direction  of  transmission  is 
likely  to  move  rapidly  through  a  wide  horizontal  angle,  antenna  gain  is  less  important,  and 
a  directional  antenna,  unless  it  has  pronounced  nulls,  is  frequently  found  satisfactory  be- 
cause the  unfavorable  ship  position  is  offset  by  the  short  distance.  Otherwise,  a  simple 
antenna  is  provided  for  this  purpose. 

BIBLIOGRAPHY 

Articles  in  Proc.  I.R.E;  Elec.  Rev.  (London) ;  Elec.  Communication;  Bett  Syst.  Tech.  J.;  Trans.  AJ.I5.E.; 
Elect.  Engg.;  Proc.  I.E.E.  (EngL);  Electrician;  Wireless  World;  Bett  Lab.  Rec.;  Rev.  gen. 
Ann.  des  -pastes. 


TELEPHONE  LINES— TRANSMISSION 
CONSIDERATIONS 

13.  TYPES  OF  PLANT 

SUBSCRIBER  LINES  (LOOPS)  include  all  types  of  outside  plant  facilities  needed  to 
connect  the  subscriber  station  telephones  with  their  local  central  office.  Such  facilities 
may  consist  of  cable,  either  aerial  or  underground,  open  (bare)  wire,  carrier  or  radio  chan- 
nels, or  a  combination  thereof.  Generally,  cable  is  used  in  urban  areas  as  a  means  of 
serving  local  subscriber  stations;  where  small  urban  open-wire  plants  are  still  in  operation 
they  are  rapidly  being  replaced.  In  rural  areas,  open-wire  lines  are  still  in  general  use 
except  in  congested  sections,  since  the  fewer  lines  and  longer  distances  characteristic  of 
rural  areas  make  the  open  wire  more  economical.  As  stated  above,  however,  it  is  an- 
ticipated that  radio  facilities  can  be  made  available,  commercially,  to  serve  distant  or  rel- 
atively inaccessible  farms  where  the  costs  of  providing  the  usual  telephone  wire  facilities  would 
be  excessive. 

TOLL  LINES,  as  generally  defined,  consist  of  various  types  of  outside  plant  facilities 
employed  to  provide  toll  circuits  between  toll  centers  (TC).  Those  line  facilities  connect- 
ing TC  and  tributary  offices  are  considered  part  of  the  TC  plant.  These  latter  offices,  in 
the  general  meaning  of  the  word  "tributary,"  are  small  offices  (in  territory  adjacent  to  the 
TC)  connected  to  the  TC  by  one  or  more  tributary  circuits  and  are  fully  or  partly  depend- 
ent upon  the  TC  for  the  handling  of  their  toll  traffic.  Toll  facilities  may  consist  of  cable, 
aerial  or  underground,  open  wire,  carrier  or  radio  channels,  or  a  combination  of  these. 

Subscriber  line  facilities,  known  generally  as  exchange  plant,  and  toll  line  facilities  should 
be  designed  and  constructed  to  meet,  cooperatively,  the  overall  service  objectives  which 
are  known  collectively  as  service  standards.  These  standards  are  not  fixed  for  all  time  but 
change  with  service  needs  and  advancements  in  the  art  of  communications. 

TRANSMISSION  AND  SIGNALING  are  two  fundamental  factors  to  consider  in  any 
telephone  system,  whether  the  connecting  facilities  between  subscribers  are  wire,  carrier, 
radio,  or  a  combination  of  two  or  more  of  these  types.  If  either  transmission  or  signaling, 
or  both,  are  not  satisfactory  for  a  given  telephone  system,  the  system  is  not  workable 
under  modern  standards  of  service. 

14.  SERVICE  KEQUIEEMENTS— TOLL 

UNIVERSAL  SERVICE  is  the  goal  toward  which  the  telephone  industry  has  been 
striving  for  many  years  and  which,  it  now  appears,  will  be  attained.  This  goal  simply 
means  that  anyone,  anywhere,  can  talk,  telephonically,  with  anyone  else,  anywhere  else, 
whether  the  connection  be  established  locally,  within  the  nation,  or  between  any  two  coun- 
tries in  the  world.  For  a  number  of  years  it  has  been  possible  to  talk  by  telephone  from 
any  point  in  the  United  States  to  any  other  point  connected  to  the  nationwide  toll  system 
and  to  many  foreign  countries.  Worldwide  service  is  being  rapidly  expanded  to  include 
those  countries  not  at  present  reached. 

TRANSMISSION  OF  SPEECH  between  two  points  requires  that  speech  (sound) 
power  from  the  talker  actuate  his  transmitter  diaphragm  and  that  the  transmitter  con- 
vert this  power  into  electrical  power,  which  travels  to  the  distant  listener's  receiver,  where 


17-70 


TELEPHONY 


it  is  reconverted  by  the  receiver  diaphragm  into  speech  (sound)  power  of  approximately 
the  same  characteristics  as  the  original  speech  power.  It  is  obvious  that  the  electrical 
power  will  diminish  as  it  travels  over  the  circuit  between  the  talker  and  listener  stations 
as  the  result  of  series  and  shunt  impedances  encountered  in  the  lines  and  equipment  which 
constitute  the  circuit.  If  the  electrical  power  reaching  the  receiver  is  diminished  to  the 
point  where  it  no  longer  drives  the  receiver  diaphragm  sufficiently  to  permit  the  listener's 
ear  to  interpret  the  intelligence  carried  by  the  resulting  sound  waves,  then  the  electrical 
transmission  loss  in  the  circuit  is  too  high  to  permit  carrying  on  a  satisfactory  telephone 
conversation.  It,  therefore,  becomes  necessary  to  limit  overall  transmission  losses  and 
consequently  the  losses  in  component  parts  of  circuits,  which  usually  consist  of  two  or 
more  sections  or  links. 

THE  GENERAL  TOLL  SWITCHING  PLAN  (Fig.  1)  as  developed  for  establishing  toll 
connections  on  a  manual  basis  provides  a  practicable  plan  for  accomplishing  universal 


"SEE 

NOTE1 

[+3] 

JO 

1 

g 

TC 


FOR  CIRCUITS  DESIGNED  TO  HANDLE  ONLY  TERMINAL  BUSINESS  THE  WORKING  NET  LOSS 
DEPENDS  UPON  TOLL  TERMINAL  LOSSES  INVOLVED  AND  UPON  OVERALL  DIRECT  STANDARDS. 
AS  A  TYPICAL  EXAMPLE,  WITH  A  DIRECT  STANDARD  OF  18  DECIBELS,  THE  LIMITING  VALUES 
OF  TERMINAL  CIRCUIT  NET  LOSSES  WOULD  BE  AS  SHOWN  BELOW  FOR  THE  ASSUMED  VAL- 
UES OF  TOLL  TERMINAL  LOSS. 


PO         1Q          Pp 


PO 


13          TC 


TC         16 


TC 


NOTES: 

1.  THE  TOLL  TERMINAL  LOSS  WOULD  BE  DETERMINED  IN  EACH  CASE  ON  THE  BASIS  OF  MEETING  THE 
TRANSMISSION  REQUIREMENTS  IN  THE  MOST  ECONOMICAL  AND  SATISFACTORY  MANNER,  FOR  EX- 
AMPLE, IN  MEETING  THE  10  DECIBEL  UMJT  FOR  OUTLET  TERMINAL  LOSS.   THE  VALUES  SHOWN 
ABOVE  ARE   TYHCAL  BUT  THE  ECONOMICAL  TOLL  TERMINAL  LOSS  IS  EXPECTED  TO  VARY  CONSID- 
ERABLY IN  INDIVIDUAL  CASES. 

2.  PAD  VALUES  DEPEND  ON  NOISE  AND  CROSSTALK  CONDITIONS,  ON  LIMITING  TOLL  TERMINAL  LOS- 
SES AND  ALSO,  IN  THE  CASE  OF  INTERMEDIATE  LINKS,  ON  ECHO  MARGIN  REQUIREMENTS. 


STANDARDS 

2 -LINK  THRU  PO- 
2 -LINK  THRU  TC 

3 -LINK • 

4 -LINK 

5  -  LINK 23 

OUTLET  TERMINAL  LOSS — 10 


TC 


CODE 

TERMINATING  TOLL 
CENTER 
PO      PRIMARY  OUTLET 
RC      REGIONAL  CENTER 
•VvV  SWITCHING  PAD 

INDICATES  TYPICAL 
TOLL  TERMINAL  LOSS 


(6)  MINIMUM  WORKING 
ECHO  NET  LOSS  (AS- 
SUMES NO  TRANSMIS- 
SION IMPAIRMENTS) 

9    EFFECTIVE  WORKING 
NET  LOSS -VIA  CONDITION 

[+3]  ECHO  MARGIN 


FIG.  1.    Typical  Example  of  the  General  Toll  Switching  Plan,  Showing  Limiting  Toll  Circuit  Losses 
_  (Courtesy  Bell  System) 

service  in  the  United  States  and  throughout  the  world,  with  the  development  and  ex- 
pansion of  telephone  systems  in  other  countries. 

From  Table  1  it  will  be  noted  that  any  two  subscriber  telephones  which  have  access  to 
the  nationwide  toll  network  can  be  connected  together,  using  not  more  than  five  toll  cir- 
cuit links  and  four  switches.  It  is  assumed  that  the  telephones  are  located  at  their  re- 
spective toll  center  (TC)  points,  i.e.,  not  at  tributary  points,  which  would  necessitate 
using  a  tributary  trunk  to  reach  the  respective  TC  office. 

The  plan  provides  for  eight  regional  centers  (RC},  Atlanta,  Chicago,  Dallas,  Denver, 
Los  Angeles,  New  York,  San  Francisco,  and  St.  Louis,  each  being  strategically  located 
within  the  United  States,  to  serve  as  toll  switching  centers  of  the  first  order.  Each  of  these 
centers  is  connected  by  direct,  high-grade  toll  circuits  to  each  of  the  other  centers.  Within 
each  RC  area  are  a  number  of  important  toll  centers  known  as  primary  outlets  (PO),  each 


SERVICE  REQUIREMENTS — TOLL 


17-71 


being  connected  by  direct,  high-grade  toll  circuits  to  its  own  RC,  other  RCs  and  other 
POs,  as  required  to  best  handle  the  traffic.  Finally,  each  PO  serves  directly  all  the  toll 
centers  (TC}  within  its  area,  and  the  TCs  serve  the  subscribers  within  their  local  areas, 
either  directly  or  through  their  tributary  offices.  Thus,  the  plan  provides  for  a  con- 
centration of  the  toll  traffic  at  the  various  toll  centers  which  have  access  to  or  are  ac- 
cessible from  any  part  of 

the     nationwide     system  Table  1.    Overall  Standards 

(including  Bell  and  Inde- 
pendent) through  direct 
or  switched  connections. 

Table  1  shows  overall 
standards  and  number  of 
links '  and  switches,  and 
Table  2  shows  assigned 
losses  for  the  different  toll 
links,  under  the  present 
general  toll  switching 
plan.  *  The  letter  codes  are  defined  in  Fig.  1. 

In     general,     four-wire 

circuits  (a  separate  path  for  each  direction  of  transmission)  or  carrier  channels  are  em- 
ployed for  the  long-haul,  intermediate  toll  links  because  of  their  better  performance  at 
low  losses  than  that  of  two-wire  circuits.  Two-wire  circuits  are  generally  used  for  the 
shorter  end  links  and  toll  trunks. 

Table  2.    Allowable  Toll  Link  (Circuit)  Losses 


ToH  Connection  * 
(for  switched  traffic) 

Overall 
Standard, 
decibels 

Number  of 
Circuit 
Links 

Number  of 
of 
Switches 

Direct   . 

17-20 

1 

0 

TC-TOTC  

22 

2 

I 

TC-PO-PO-TC  

21 

3 

2 

TOPO-RC-TC 

21 

3 

2 

TC-PO-RC-RC-TC  

22 

4 

3 

TC-PO-RC-RC-PO-TC  

23 

5 

4 

Toll  Link 
(for  switched  traffic) 

Effective 
Working 
Net  Loss, 
decibels 

Minimum  Working  Echo 
Net  Loss,  decibels 
(assumes  no  transmission 
impairments) 

Echo 
Margin, 
decibels 

TC-PO  (end  link)          

IO-(TTL)  * 

7-(TTL) 

-r3 

PO-PO  (intermediate  link)  

1 

3 

-2 

PO-RC  (intermediate  link)  
RC-RC  (intermediate  link)  
PO  terminal  loss  

I 

1 

lot 

3 
3 

-2 
-2 

*  This  value  depends  upon  the  most  economical  and  practicable  toll  terminal  loss  for  each  individual 
toll  center  which  will  meet  the  required  primary  outlet  (PO}  terminal  loss  of  10  db. 

t  This  value  results  from  taking  one-half  of  the  loss  (20  db)  for  a  tw>4ink  connection  through  a 
gain  center  (PO  or  RC). 

The  PO  terminal  loss  of  10  db  is  fixed,  unless  changed  under  the  plan.  This  loss  may  be 
allocated  to  the  TC-PO  circuit  and  the  toll  terminal  loss  as  required.  Toll  terminal  losses 
(TTL)  vary  from  about  0  to  5  db. 

TOLL  CIRCUIT  OPERATING  REQUIREMENTS.  The  minimum  working  net  loss 
(MWNL}  of  a  toll  circuit  is  the  lowest  net  loss  that  may  be  assigned  that  will  satisfy  the 
design  objectives  imposed  by  singing,  echo,  crosstalk,  and  noise,  when  subject  to  maximum 
negative  transmission  variations.  

Tlie  minimum  working  echo  net  loss  (MWENL)  is  the  lowest  1000-cycle  net  loss  which 
can  be  assigned  so  that  a  circuit  will  satisfy  the  echo  objectives,  including  the  assigned 
echo  margin.  If  the  loss  at  which  a  circuit  is  operated  is  greater  than  the  loss  required 
to  offset  the  echoes  arising  from  the  circulating  current  paths  within  the  circuit,  a  positive 
echo  margin  is  said  to  result.  If  the  reverse  is  the  case,  a  negative  margin  will  be  intro- 
duced. In  order  to  operate  the  plant  at  lowest  over-all  losses  on  built-up  or  switched 
connections  consisting  of  two  or  more  TvnVs,  positive  echo  margins  have  been  assigned  to 
some  classes  of  circuits  and  negative  margins  have  been  assigned  to  others.  Switching 
arrangements  are  so  designed  that  negative  margins  will  be  offset  in  any  connection. 

Echoes  are  the  result  of  imperfect  balances  in  toll  circuits  equipped  with  telephone 
repeaters  and  four-wire  terminating  sets.  The  two-wire  circuits  are  necessarily  converted 
to  four-wire  circuits  at  each  repeater,  and  the  four-wire  circuits  require  four-wire  terminat- 
ing sets  at  their  terminals  to  convert  the  four-wire  circuit  to  two-wire  before  extending 
it  to  the  toll  switchboard.  At  certain  important  switching  offices,  four-wire  switching 
on  a  mechanical  basis  may  be  applied.  At  the  points  of  conversion,  a  balancing  network 
terminates  one  branch  of  the  hybrid  coil,  and  the  opposite  branch  of  this  coil  is  connected 
to  the  outgoing  or  incoming  toll  circuit  or  the  extension  to  the  switchboard.  It  is  not 
practicable  to  match  the  impedance  of  the  toll  circuit  exactly  or  the  extension  with  the 


17-72 


TELEPHONY 


impedance  of  the  network.  Thus,  part  of  the  voice  currents  are  transferred  across  the 
hybrid  bridge  into  the  repeater  inputs  in  varying  degrees  at  each  repeater  or  terminating 
set  on  the  circuit  instead  of  dividing  equally  between  the  outgoing  line  and  its  balancing 
network.  These  currents  which  enter  the  repeater  inputs  will  be  amplified  and  travel 
back  to  the  talker  with  some  delay,  so  that  he  hears  his  own  words  (in  reduced  volume) 

coming  back  to  him  an  instant  after  he  has 
spoken  them.  The  listener  may  also  be  af- 
fected. As  the  delay  and  return  volume  in- 
crease, the  annoyance  becomes  greater  and 
may  reach  the  point  at  which  the  conversation 
is  not  satisfactory.  Figure  2  shows  minimum 
working  net  losses  for  terminal  grade  circuits 
as  limited  by  echoes,  versus  typical  lengths  of 
toll  cable  without  echo  suppressors  and  with 
anti-sidetone  sets. 

To  overcome  echo  difficulties,  repeater  gains 
must  be  properly  assigned  and  regulated.  Also, 
an  echo  suppressor  has  been  developed  which, 
by  voice  current  action,  causes  a  high  loss  to 
be  introduced  in  one  side  of  the  four-wire  cir- 
cuit while  speech  is  being  transmitted  on  the 
other  side.  A  simple  schematic  of  this  device 
is  shown  in  Fig.  3.  Generally,  two-wire  cir- 
cuits are  relatively  short  and  do  not  require 


f 


1/7 


v/ 


EVEN  HYBRID 


EVEN 

TRANSMISSION 
PATH 


o      too    200    300    4oo    soo    eoo    700  echo  suppression. 

DISTANCE  IN  MILES  The    minimum   working    singing    net   loss 

FIG.  2.    Minimum  Working  Net  Loss  of  Ter-    (M"WSNL)  is  the  lowest  net  loss  assignable  to 

minal  Grade  Circuits  as  Limited  by  Echoes,    a  circuit  which  will  meet  singing  requirements 

^ut^lf  sSLSS?  £F™2«h"/£t£  **  indicated  above  under  "echoes,"  the  unl 

sidetone  Sets  (Courtesy  Bell  System)  balances  existing  in  practice  at  the  hybrid  coils 

of  telephone  repeaters  and  four-wire  terminat- 
ing sets  cause  part  of  the  outgoing  energy  from  one  branch  of  the  four-wire  circuit  to 
pass  through  the  hybrid  coil  bridge  points  to  the  opposite  transmitting  branch  of  the 
four-wire  circuit.  At  each  hybrid  coil  this  action  occurs,  so  that,  whether  the  circuit 
consists  of  only  a  two-wire  repeater  with  terminating  lines  or  a  long  four-wire  circuit  with 
several  intermediate  repeaters  and  a  four-wire  terminating  set  at  each  end  of  the  circuit, 
a  circulating  current  is  established,  provided  that  the  net  circuit  gains  exceed  the  net 
circuit  losses  in  the  circulating  current 

circuit  and  the  phase  change  in  the  cir-  ''OWV 

culating  current  is  a  multiple  of  360°. 

It  is  thus  necessary  in  designing  cir- 
cuits, particularly  two-wire,  to  limit 
these  circulating  currents  by  assigning 
repeater  gains,  so  that,  for  an  average 
circuit  net  loss,  the  most  critical  two- 
wire  repeater  for  95  per  cent  of  the 
connections  will  have  losses  which  total 
at  least  10  db  more  than  the  gains  in 
the  two  directions  of  transmission.  For 
the  short  terminal  circuits  with  one  or 
two  repeaters,  an  8-db  singing  margin 
will  usually  be  satisfactory. 

Figure  4  shows  minimum  working 
net  losses,  as  limited  by  singing,  versus 
typical  toll  cable  circuit  lengths  for 
specified  conditions  of  repeater  spacing 
and  singing  points. 

Tne  TniTtiTntiTn  working  crosstalk  net 
loss  (MWXNL)  is  the  lowest  net  loss 
assignable  to  a  circuit  which  will  satisfy 


p— i — nsw^ — 
•=•     L— W\ 

EVEN  RELAY 


FIG.  3.   Schematic  of  Echo  Suppressor  Circuit  (Cour- 
tesy  Bell  System) 


crosstalk  requirements  under  all  operating  conditions.  Crosstalk  is  the  electric  and  mag- 
netic transfer  of  speech  or  similar  currents  from  one  telephone  message  circuit  to  another. 
It  may  or  may  not  be  intelligible,  but  when  it  is  composed  of  confused  noise  from  several 
sources  it  is  known  as  babble. 

Crosstalk  usually  results  from  cumulative,  slight  unbalances  between  circuits  or  high 


SERVICE  REQUIREMENTS — TOLL 


17-73 


energy  level  differences  acting  through  close  couplings  in  cable  or  open  wire.     The  trans- 
ferred energy  is  amplified  wherever  telephone  repeaters  are  present. 

Two  types^  of  crosstalk,  near  end  and  far  end,  develop  between  circuits.     The  first  type 
travels  to  a  listener  on  one  circuit  in  a  direction  opposite  to  the  transmission  from  a  talker 
on  another  circuit;  the  latter  type  travels  to  the  listener  on  one  circuit  in  the  same  direction 
as   the  transmission  from  a 
talker    on    another    circuit. 
Near-end  crosstalk  occurs  in 
wire  but  not  in  properly  ar- 
ranged  carrier   circuits    (ex- 
cept W.E.  Co.  G-type) ;  far- 
end  crosstalk  appears  in  both 
wire  and  carrier  circuits. 

The  effect  of  crosstalk  on 
subscriber  conversations  de- 
pends not  only  on  the  actual 
volume  of  crosstalk  heard 
but  also  on  circuit  and  room 
noise  present,  circuit  losses, 
and  personal  reactions. 

Crosstalk  is  controlled  pri- 
marily by  avoiding  excessive 
energy  level  differences  and 
couplings ^  between  adjacent 
parallel  circuits.  Techniques 
have  been  developed  to  limit 
both  level  differences  (by 


3 


100     200     300     400 


500 
Miles 


600     700     800     900    1000 


FIG.  4.    Minimum  Working  Net  Loss  as  Limited  by  fc 
Typical  Toll  Cable  Circuit  Lengths  (Courtesy 


;  versus 
tern) 

proper  regulation  of  repeaters  and  other  amplifiers)  and  high  couplings  and  to  employ 
different  frequency  bands  in  controlling  crosstalk. 

Figure  5^  shows  minimum  working  net  losses,  as  limited  by  crosstalk,  versus  typical  toft 
cable  circuit  lengths. 

Crosstalk  values  have  generally  been  expressed  in  terms  of  crosstalk  units,  which  are 
defined  as^one  million  ^lO6)  times  the  ratio  of  the  crosstalk  current  or  voltage  at  the  ob- 
serving point  on  the  disturbed  circuit  to  the  current  or  voltage  at  the  sending  point  on  the 
disturbing  circuit  (assuming  equal  impedances  at  these  two  points).  If  the  impedances 
are  not  equal,  the  square  root  of  the  power  ratio  may  be  used  in  place  of  the  current  or  volt- 
age ratio.  With  the  development  of  visual  indicating  apparatus  for  measuring  crosstalk 
and  noise,  crosstalk  measurements  have  more  generally  been  made  in  terms  of  crosstalk 
coupling  loss-db^  which  means  the  net  transmission  loss  between  the  sending  point  on  the 
disturbing  circuit  and  the  receiving  point  on  the  disturbed  circuit,  it  being  understood  that 

the  higher  the  measuring  set  reading  (loss), 
the  less  the  actual  coupling.  More  recently, 
the  term  db  above  reference  coupling-dbx  has 
come  into  use.  This  term  means  the  cou- 
pling in  decibels  above  reference  coupling, 
and  reference  coupling  means  the  coupling 
which  would  be  required  to  give  a  reading 
of  0  dba  on  a  W.E.  Co.  2B  noise-measuring 
set  connected  to  the  disturbed  circuit  when  a 
test  tone  of  90  dba  (using  the  same  weighting 
as  on  the  disturbed  circuit)  is  impressed  on 
the  disturbing  circuit. 

The  2B  set  is  designed  to  measure  cross- 
talk and  noise  volumes  or  couplings  (as  well 
as  other  quantities)  in  decibel  values.    These 
~T4  values  can  be  adjusted  to  a.  common  basis 
for  different  types  of  lines  and  telephone 
FIG   5.    Minimum  Working  Net  Loss  as  Limited  receivers  so  that  a  given  adjusted  value,  des- 
by  <**«*  C—'ignated  dba,  will  mean  the  same  interfering 

effect  to  the  ear,  regardless  of  the  type  of 


8000T 

6000 
5000 
^4000 
=  3000 

s- 

^2000 

1 
^  1000 

i 

i 

i 

H4 

4-2 

5( 

t  W 

re) 

— 

y 

^ 

a    800 
£    600 
0    500 
400 
300 

200 
100 

HF 

•^ 

; 

^ 

s 

H4 

4-2 

5(2 

>  W 

re) 

A 

/ 

/ 

\\ 

/ 

/ 

/ 

I 

4 

38J 

-5C 

an 

i  H 

88- 

50 

2          4          6  8         10        12 

Minimum  Working  Net  Loss  (Decibels) 


line  or  subscriber  set,  affected  by  the  crosstalk  or  noise  being  measured. 

Figure  6  shows  the  relation  between*  the  terms  crosstalk  units  (cu),  crosstalk  coupling 
loss,  db,  and  crosstalk  coupling,  dbx. 

The  present  design  requirements  for  crosstalk  limitations  in  circuits  are  taking  into 
account  the  wide  reactions  of  different  people  to  different  amounts  of  crosstalk,  the  vari- 


17-74 


TELEPHONY 


ation  in  crosstalk  volumes  due  to  the  variation  in  speech  power  of  different  talkers,  the 
action  of  room  and  line  noise  on  crosstalk  effects,  the  intelligibility  of  crosstalk,  costs  in- 
volved in  its  control,  and  attainment  of  a  good  balance  in  judging  the  importance  of  the 

various  factors  entering  into  circuit  design. 

CROSSTALK  COUPLING  LOSS  IN  DECIBELS  Present  practices  indicate  that  it  is  inadvisable 
too      90       so       70      60      50    _40   to  perniit  crosstalk  couplings  in  excess  of  30 

dbx  (equivalent  to  60  db  crosstalk  coupling  loss 
or  to  1000  crosstalk  units),  as  measured  from 
the  transmitting  to  the  receiving  switchboard 
for  a  single  disturber  circuit. 

Noise  transmission  impairments  which  may 
exist  in  toll  circuits  due  to  power  line  induc- 
tion, noise  generated  within  telephone  systems, 
or  circuit  irregularities  must  be  limited  to 
avoid  transmission  penalties  in  circuit  opera- 
tion. Noise  reaching  the  subscriber's  ear 
through  his  telephone  receiver  is,  in  fact, 
equivalent  to  adding  loss  to  the  circuit.  Table 
3  shows  how  these  penalties  are  evaluated  in 
terms  of  noise  levels-dba. 

Reference  noise  (RN)  is  used  as  a  base  in 
the  calculation  of  circuit  noise  in  terms  of  dec- 
ibel penalties,  as  given  in  Table  3.  Reference 
noise  is  registered  as  0  dba  on  a  2B  noise- 
measuring  set  when  the  input  into  this  set  is 
10  ~12  watt  of  1000-cycle  power  (line  weight- 
ing). UN  is  equivalent,  when  measured  at 
the  terminals  of  a  600-ohm  line  (with  line 
weighting),  to  7 noise  units;  if  measured  across 
the  terminals  of  a  W.E.  Co.  No.  144  receiver, 


IO,OOO 
8,000 
6,000 

4.OOO 
2,000 

3   1,000 
.     800 
^      600 

3      400 

} 

•>      100 
80 
fin 

/ 

/ 

/ 

/ 

f 

/ 

/ 

/ 

/ 

/ 

/ 

/ 

/ 

40 

/ 

20 

/ 

10 

/ 

-10        o        TO      20      30      40      so   it  is  equivalent  to  14  noise  units. 

CROSSTALK  COUPLING  ,  DBX  Circuit  noise  of  29  db  above  RN  (200  noise 

FIG.  6.  Chart  showing  Relation  between  Cross-  units)  Or  less  in  a  600-ohm  line  is  not  con- 
talk  Units,  cu;  Crosstalk  Coupling  Loss  in  sidered  to  offer  any  appreciable  noise  impair- 
Decibels;  and  Decibels  above  Reference  ^  j  j.-  i  ±  *  \.  ±  i 

Coupling,  dbx  (Courtesy  BeE  System)  ment  to  a  conversation,  but  for  about  each 

3-db  increase  in  noise  level  above  29  db  the 
impairment  increases  1  db,  which  must  be  included  in  the  overall  circuit  loss. 

Remedial  measures  have  been  perfected  for  controlling  most  types  of  noise  to  avoid 
penalties. 

Distortion  transmission  impairment  (DTI)  to  conversations  results  from  a  restricted  or 
modified  transmission  of  the  full  voice-frequency  band  necessary  for  clear,  understandable 
speech.  Such  restriction  or  modifica- 
tion may  be  due  to  a  low  cutoff  fre- 
quency of  certain  types  of  loaded  line 
facilities  and  line  apparatus.  The  older 
H172-63  loaded  cable  facilities  and  cer- 
tain early  types  of  telephone  repeaters, 
carrier  systems,  and  filters  give  distor- 
tion impairments.  The  latest  types  of 
loaded  cable  facilities,  such  as  H44-25 
and  H  and  B88-50,  and  the  latest-type 
standard  repeater  and  carrier  systems 
are  considered  to  offer  no  appreciable 
distortion  for  the  usual  lengths  em- 
ployed. Figure  7  and  Table  4  show 
distortion  impairments  for  different 
facilities  with  FIA-AST  subscriber  sets 
and  H88  switching  trunks,  as  used  in 
the  Bell  System. 

Volume  transmission  losses  in  toll 
circuits  will  vary  with  changes  in  tem- 
perature and,  in  addition,  for  open-wire 


IN  DECIBELS 

\ 

EFFECTIVE  CUTOFF  FREQUENCY  IS 
THAT  FREQUENCY  AT  WHICH  THE 
OVERALL  NET  LOSS  OF  THE  TOLL 
CIRCUIT  IS  1O  DECIBELS  GREATER 
THAN  THAT  AT  1000  CYCLES, 

\ 

\ 

\ 

N 

X 

s^ 

\ 

**^ 

•^^ 

***** 

L8      2.0      2.2      2.4      2.6      2.8      3.0      3.2      3.4      3.6 
0  EFFECTIVE  CUTOFF  FREQUENCY  OF  TOLL  CIRCUIT 

IN  KILOCYCLES  PER  SECOND 

FIG.  7.  Diagram  Showing  Distortion  Transmission 
Impairment  for  Different  Facilities  with  FIA-AST 
Subscriber  Set  and  H-88  Switching  Trunk,  versus  Ef- 
fective Cutoff  Frequency  of  Toll  Circuit  (Courtesy  Bell 
System) 


facilities  with  such  conditions  as  rain,  sleet,  and  snow.  These  transmission  variations 
are  different  for  aerial  and  underground  cable  and  open  wire  and  also  change  with  fre- 
quency. 


SERVICE  REQUIREMENTS — TOLL 


17-75 


Table  3.    Noise  Transmission  Impairments  Corresponding  to  Uoise  Magnitudes  Measured 
or  Computed  at  Various  Points  in  the  Transmission  Circuit 


Type  of  circuit  < 

Toll  circuits 

Toll  connecting, 
tandem  and  in- 
teroffice trunks 

Local  loops 

Point  of  i 
estima1 

neasurement  or  J 
e               .          1 

Receiving  toll 
switchboards 

Local  office 

Subset 
terminals 
of  loop 

Across    receiver 
terminals 

I 

Impedan 
device 

ce  of  Measuring  f 
1 

600    ohms,    termi- 
nating, 6000  ohms 
bridging  * 

600    ohms,    termi- 
nating, 6000  ohms 
bridging  * 

600  ohms 
terminating 

Approximately 
2000  ohms 

Noise 
Rating 

Noise 
Transmission 
Impairment, 
decibels 

Noise  Magnitudes  in  dba 

Line  Weighting 

Line  Weighting 

Line 
Weighting 

Receiver 
Weighting 

NO 

Nl 
N2 
N3 
N4 
N5 
N6 
N7 

0 
1 
2 
3 
4 
5 
6 
7 

0    -29 
29.1-32 
32.  1-35 
35.1-38 
38.  1-40 
40.1-42 
42.  1-43 
Over  43 

0    -26 
26.  -29 
29.  -32 
32.  -35 
35.  -37 
37.  -39 
39.  -40 
Over  40 

0     -20 
20.   -23 
23.  -26 
26.  -29 
29.  -31 
31.  -33 
33.  -34 
Over  34 

0-17 
17.  -20 
20.  -23 
23.  -26 
26.  -28 
28.  -30 
30.  -31 
Over  31 

*  When  bridging  measurements  are  made  on  a  line,  600-ohm  impedance  is  assumed  each  way  from 
the  bridging  point.  Under  these  conditions  (a)  with  144  rec-  line  weighting,  a  correction  f  of  +11  db 
is  added  to  2B  set  readings  and  (6)  with  FIA  line  weighting,  a  correction  f  of  about  -f  18  db  is  added 
to  2B  set  readings,  in  order  to  express  readings  in  dba.  If  impedances  each  way  from  bridging  point 
are  not  600  ohms,  correct  2B  set  readings  as  follows: 

Each  impedance  300  ohms,  correction  is  +3  db. 

Each  impedance  400  ohms,  correction  is  +1-8  db. 

Each  impedance  900  ohms,  correction  is  — 1.8  db. 

Each  impedance  1200  ohms,  correction  is  —3.0  db. 

Each  impedance  2000  ohms,  correction  is  —5.2  db. 

t  The  corrections  thus  indicated  should  be  added  before  entering  Table  3. 

Present  Bell  System  practices  provide  for  maintaining  toll  circuits  of  minimum  to  max- 
imum lengths  within  the  following  deviations  from  the  specified  value:  voice-frequency 
(VF)  cable,  ±1.0  to  ±4.0  db;  K-carrier,  ±2.0  db;  VF  open  wire,  ±1.0  to  ±3.0  db; 
open-wire  carrier,  ±2.0  db;  and  various  combinations  and  lengths  of  facilities,  ±1.0  to 
±4.5  db. 

Such  limitation  is  provided  for  long  VF  cable  circuits  by  devices  known  as  automatic 
transmission  regulators  spaced  at  proper  intervals  along  the  circuit  to  automatically  acid 
or  reduce  gain  in  the  circuit  as  required  to  maintain  the  specified  volume  limits.  Figure 
8  shows  a  pilot  wire  transmission  regulator  circuit,  with  its  pilot  wire  cable  pair  in  the  same 
cable  as  the  regulated  cable  circuits.  Long  cable  circuits  have  a  regulating  repeater  (in 
place  of  the  regular  repeater)  about  every  150  miles.  These  repeaters  are  oontroEed  at 
each  point  by  a  master  regulator,  in  accordance  with  temperature  change  in  the  pilot  wire, 
which  causes  the  repeater  gains  to  vary  above  or  below  normal  setting  in  steps  varying 
from  1/4  to  1  db  over  a  range  varying  from  2.75  to  19  db  as  required  to  maintain  normal 
level.  No  system  of  automatic  regulation  has  seemed  necessary  for  use  with  open-wire 
voice-frequency  facilities. 

Since  in  cable  circuits  the  attenuation-frequency  curve  is  appreciably  different  at  dif- 
ferent temperatures,  it  is  necessary  to  correct  for  this  difference,  known  as  twist,  for  high- 
frequency  carrier  systems,  such  as  the  K-system.  The  twist  effect  in  a  100-mile  aerial 
toll  cable  is  shown  in  Fig.  9.  Twist  correcting  circuits,  as  shown  in  Fig.  10,  are  located 
in  long  cable  circuits  about  every  100  miles  for  aerial  and  200  miles  for  underground  cable. 

For  open-wire  carrier  systems,  pilot  channel  regulator  equipment  is  incorporated  in 
the  carrier  terminal  and  repeater  design  to  maintain  transmission  levels. 

The  effective  transmission:  loss  of  a  toll  circuit  is  equal  to  the  1000-cyde  loss  plus  any 
noise  or  distortion  transmission  impairments,  all  expressed  in  decibels. 

The  overall  effective  equivalent  of  a  complete  toll  connection  from  subscriber  to  sub- 
scriber is  the  sum  of  the  effective  transmission  loss  of  the  toll  circuit  or  circuits  and  the  toU 
terminal  losses  at  the  terminating  toll  centers. 


17-76 


TELEPHONY 


Table  4.     Distortion  Impairments 


Voice  Frequency 
Facility 

2-Wire 
or 
4-Wire 

Filter 
(Note  4) 

Length  in  Miles  for  Various  Impairments 

-1  db 

Odb 

-Hdb 

4-2  db 

+3db 

+4db 

+5db 

+6db 

H245-S 

2-W 
2-W 

No  repeater 

A 

0-10 

10-40 
0-7 

40-70 
7-30 

70-100 
30-50 

H155-P 

2-W 
2-W 

No  repeater 
A 

0-20 

20-60 
0-10 

60-100 
10-40 

40-50 

H174-S,H172-S 

2-W 
2-W 
4-W 
4-W 

No  repeater 
B 
B 

A 

0-35 
0-15 

35-70 
15-50 

70-100 
50-90 
0-300 

90-160 

160-270 
Over  300 

m  E106-P 

2-W 
2-W 
4-W 
4-W 

No  repeater 
B 
B 
A 

0-60. 
0-25 

60-110 
25-60 

60-150 
0-300 

150-220 

220-300 
Over  300 

H63-P 

2-W 
2-W 
4-W 
4-W 

No  repeater 
C 
C 
B 

Any 

0-300 

0-75 
300-700 

75-180 
Over  700 

180-450 

BorE8S-50,SorP 

2-W 
2-W 
2-W 
4-W 
4-W 

No  repeater 
D 
C 
D 
C 

Any 
0-150 

0-200 

150-450 

0-100 

Any 

100-250 

250-400 

Over  400 

H44-25,  S  or  P 

2-W 
2-W 
2-W 
4-W 
4-W 

No  repeater 
D 
C 
D 
C 

Any 
0-150 

0-800 

150-450 
0-1000 

0-100 

Over 
1000 

100-320 

Over  320 

N.L.  open-wire  side 
with  3KC  carrier 
line  filters 

2-W 
2-W 
2-W 

No  repeater 
1059B 
C 

Any 

0-800 
0-300 

Over  800 
Over  300 

N.L.  open-wire 
phan.  or  sides 
with  5  kc  or  no 
carrier  line  filters 

2-W 
2-W 
2-W 
2-W 

No  repeater 
D 
1059B 
C 

Any 
Any 

0-800 
0-300 

Over  800 
Over  300 

Type  of  Carrier  Frequency  Circuit 

Maximum  Number  of  Links  for  Various  Impairments 

-Idb 

Odb 

-Hdb 

+2db 

+3db 

+4db 

+9db 

C2,  C3,  C4—  Manual  regula 

tion 

1 

2 

5 

C2,  C3  —  Automatic  regulati 

on 

1 

2 

4 

Over  4 

C4—  Automatic  reg 

ulation  . 

2 

5 

C5 

Any 

D        

1 

2 

EB     

Any 

GI  .    ...                

2 

HI  —  No  repeaters  

1 

3 

J  

Any 

K  

Any 

L  

Any 

Notes: 

1.  Impairments  are  referred  to  a  distortionless  toll  circuit  containing  a  250-3000  cycle  band-pass  filter  having  square- 
cutoffs. 

2.  Impairments  are  substantially  independent  of  gage. 

3.  Impairments  are  substantially  independent  of  type  of  line  repeater,  provided  standard  equalization  is  employed.. 

4.  A  refers  to  13A  or  32  A  filters;  B,  to  13B.32B  or  128B  filters;  C,  to  13C,  32C  or  12SC  filters;  D,  to  D93985,  32C 
modified  per  KS-4165  (D  160523),  or  128 A  filters.   1059B  filter  is  associated  with  the  high  level  22-type  repeater 
(104-D  tubes). 


SERVICE  REQUIREMENTS  —  TOLL 


17-77 


REGULATOR  SECTION 


c 


SHORT  SECTION 


1 


PILOT  WIRE  CABLE  PAIR                       LONG 

SECTION 

/ 

1            ^ 

.1- 

n    r 

1 

GALVANOMETER 
MECHANISM 

TO  NO.2 
REGULATOR 

! 

V 

XJx 

LJ 

MASTER 
RELAYS 


W-E 


_J 

> 

< 

r 

> 

, 

CABLE 
FIG.  8. 


-------------  REGULATOR  OFFICE  -------------  * 

Pilot  Wire  Transmission  Regulator  Circuit  (Courtesy  BeD  System) 


The  desired  overall  transmission  loss  having  been  apportioned  to  the  various  compo- 
nent parts  of  toll  connections,  as  shown  under  the  plan,  the  pattern  for  the  engineering  of 
toll  facilities  is  thus  fixed.  The  toll  ter- 
minal loss  as  determined  for  each  toll 
center  is  defined  as  the  average  (one-half 
of  the  sum)  of  the  effective  transmitting 
and  receiving  losses  (see  article  15)  from 
the  toll  circuit  termination  to  (and  in- 
cluding the  efficiency  of)  the  subscriber 
station  apparatus.  With  this  loss  deter- 
mined for  a  given  toll  center,  the  toll 
switching  trunks  and  subscriber  loops 
(exchange  plant)  must  be  engineered  to 
meet  this  requirement,  although  ex- 
change plant  engineering  also  is  subject 
to  exchange  standards. 

TOLL  CIRCUIT  LINE-UP  PRO- 
CEDURE consists  of  adjusting  the  op- 
erating gains  of  voice  frequency  and  carrier  repeaters,  carrier  system  terminals,  and  other 
associated  apparatus,  such  as  switching  pads,  equalizers,  attenuators,  and  other  devices 
necessary  for  proper  operation  of  the  toll  circuits. 


£?  DEVIATION  PROM  55*  F  NET  LOSS 
Q  IN  OECIBEUS 
'  ^LOSS  GAIN-*., 
•  ,.*>.  w  0  w  * 

\ 

\ 

^ 

TEMP.   (N 

Jr\ 

3SGRS 

no  . 

55 

ES  F 

,-~~ 

/ 

^ 

—  •-, 

*-—  .. 

-^ 

•—- 

-^ 

/ 

5        IS202S3O354O455055W 
FREQUENCY  9*  KILOCYCLES  PER  SECOND 

Twist  Effect  in  100-mile  Aerial  Cable  Circxi 
(Courtesy  Bell  System) 

17-78 


TELEPHONY 

INTERSTAGE  NETWORK 


FIG.  10.     Twist  Correcting  Circuit  for  Type  K  Carrier  System  (Courtesy  Bell  System) 

From  the  general  toll  switching  plan  (Fig.  1)  it  is  evident  that  intermediate  toll  circuits 
cannot  be  used  for  terminal  traffic  with  the  very  low  losses  assigned  without  having  ex- 
cessive echo,  singing,  crosstalk,  and  other  troubles.  It  is  thus  necessary  to  provide 
switching  loss  pads  in  each  of  these  circuits  at  each  end  and  in  the  end  link  circuits  (PO-TC) 
at  the  primary  outlet  end,  which  pads  remain  in  these  circuits  for  terminal  traffic  and  are 
automatically  switched  out  of  the  circuits  for  via  (through)  traffic.  The  total  pad  loss 
usually  employed  is  thus  equal  to  the  difference  between  the  operating  net  loss  of  a  given 
circuit  in  its  terminal  and  via  conditions. 

In  general,  when  a  circuit  meets  the  design  objectives  in  the  via  condition  and  the  loss 
of  the  switching  pad  or  pads  satisfies  the  crosstalk  objectives  in  the  terminal  condition, 
singing  will  also  be  satisfactory;  also,  echo  will  be  within  limits  if  the  pad  loss  is  at  least 
equal  to  the  assigned  negative  echo  margin. 

Transmission  levels  on  toll  circuits  require  careful  coordination  with  other  adjacent 
or  nearby  circuits  to  prevent  excessive  crosstalk.  Also  levels  should  be  relatively  high 
(within  the  capabilities  of  the  associated  equipment)  to  provide  suitable  signal-to-noise 
ratios.  Levels  of  about  -f-3  to  +6  db  generally  are  employed  at  the  input  to  two- wire 
lines,  and  from  +4  to  +10  db  at  the  input  to  four- wire  voice-frequency  lines.  Open-wire 
carrier  system  units  employ  about  +16  db  (input  to  the  line)  for  Bell-owned  systems. 

TOLL  LINE  SIGNALING  usually  employs  2CK  135-,  or  1000-cycle  signaling  systems. 
For  short  non-composited  toll  facilities,  20-cycle  signaling  is  usual;  for  the  longer  toll 
circuits  having  composite  sets  135-  or  1000-cycle  signaling  is  required.  Since  1000-cycle 
signaling  will  readily  pass  along  any  type  of  message  circuit  that  will  transmit  voice  fre- 
quencies as  such  or  in  modulated  form,  this  type  of  signaling  is  general  for  all  long  toll 
circuits.  For  crossbar  toll  operation,  as  described  in  article  3  of  this  section,  pairs  of  fre- 
quencies in  the  range  of  700  to  1700  cycles  are  sent  out  over  toll  trunks  for  pulsing  the  de- 
sired signals. 

15.  SERVICE  REQUIREMENTS— EXCHANGE 

EXCHANGE  PLANT  STANDARDS  are  based  largely  upon  giving  the  telephone-using 
public  a  convenient,  satisfactory  service  at  the  least  cost  consistent  with  protecting  the 
investment  and  employee  interests. 

The  establishing  of  exchange  plant  standards  involves  taking  into  consideration  the 
•efficiency  of  subscriber  telephone  sets  as  well  as  loop,  trunk  and  central-office  equipment 
(COE)  losses  and  signaling  ranges.  It  is  thus  necessary  to  establish  a  means  of  rating 
subscriber  loops  and  trunks  with  respect  to  a  known  transmission  standard  in  order  that 
the  capabilities  and  costs  of  the  various  types  of  equipment  and  line  facilities  that  are 
available  for  use  in  exchange  plant  may  be  compared  and  it  may  be  judged  whether  or 
in  what  respect  the  equipment  facilities  meet  the  assigned  standards  for  a  given  exchange. 

Overall  transmission  standards  are  the  upper  limits  for  the  overall  effective  station-to- 
station  transmission  and  form  the  basis  of  plant  design.  In  the  practical  application  of 
such  standards,  allowances  are  usually  deducted  for  room  noise,  and  sometimes  for  other 
impairments,  from  the  overall  standard  so  that  the  resulting  design  standard  may  be  used 
directly  in  the  design  of  the  exchange  plant. 


SERVICE  REQUIREMENTS — EXCHANGE 


17-79 


The  determination  of  the  most  desirable  transmission  standards  for  a  given  exchange  is 
a  matter  of  engineering  and  business  judgment,  based  on  a  comprehensive  view  of  local 
conditions  and  on  past  performance. 

On  the  basis  of  general  usage  of  the  W.E.  Co.  FIA-AST  or  equivalent  subscriber  sets, 
Bell  System  practices  contemplate  ultimately  overall  exchange  transmission  standards 
of  10  to  14  db  for  multiomce  exchange  area  traffic,  with  the  possibility  of  the  standards 
being  1  or  2  db  higher  for  tandem  operation.  In  single-office  areas  the  standard  is  gen- 
erally taken  as  approximately  equal  to  the  loop  limit,  considering  both  transmission  and 
signaling. 

A  working  reference  system  was  devised  some  years  ago  by  the  Bell  System  as  a  means 
of  rating  exchange  loop  and  trunk  plant.  This  system*  shown  in  Fig.  11,  includes  two 
identical  common-battery  subscriber  loops,  each  connected  through  a  24-volt  battery- 
feed  repeating  coil,  to  a  variable,  distortionless  (up  to  3000  cycles)  600-ohm  impedance 


SUBSCRIBER 
SET 


NO.  337 

TRANSMITTER 

NO.  144 

RECEIVER 

NO.  46  COIL 

STANDARD 

CONNECTION 


3 Ml.  22  GA. 
(0.082  MF 
PER  MILE) 


REPEATING  VARIABLE    REPEATING 
COIL  TRUNK  COIL 


LOOP 


SUBSCRIBER 
SET 


600W 

3000^ 

CUT-OFF 


— 

NO.  337 

3MI.22GA. 
(0.082  MF- 
PER  MILE) 

TRANSMITTER 
NO.  144 
RECEIVER 
NO.  45  COIL 

STANDARD 

— 

CONNECTION 

C)  AS  AN  ALLOWANCE 
FOR   RELAY,  OFFICE  WIRING  AND  HEAT  COILS 


LINE  NOISE  =  100  NU  IN  RECEIVER 
"TYPICAL"  ROOM  NOISE 


FIG.  11.     Working  Reference  System  for  Specification  of  Effective  Losses  (Courtesy  Bell  System) 

trunk.  The  length  of  loops  and  types  of  station  and  central-office  apparatus  are  typical 
of  conditions  and  apparatus  existing  at  the  time  the  system  was  devised  and  are  still  re- 
garded as  suitable  for  reference  purposes. 

The  working  reference  system  is  so  designed  that  it  permits  comparing  effective  trans- 
mission losses,  which  include  volume  and  distortion  losses,  line  and  room  noise,  and  side- 
tone  effects.  The  system  itself  has  an  actual  effective  transmission  loss  or  rating  of  18  db, 
based  on  7.5  db  in  the  transmitting  loop,  1.8  db  in  the  receiving  loop,  and  8.7  db  in  the 
trunk.  The  line  and  room  noises  included  in  each  loop  are  respectively  100  noise  units 
(NU)  and  room  noise  comparable  to  that  in  quiet  offices  or  fairly  noisy  residences.  This 
room  noise  is  equivalent  to  50  db  RAP  (reference  acoustic  power,  which  is  10  "^  watt  of 
sound  power  per  square  centimeter  at  the  listening  ear) . 

EFFECTIVE  TRANSMISSION  PERFORMANCE  of  exchange  telephone  plant,  as 
interpreted  in  the  United  States  and  some  other  countries,  is  evaluated  by  the  generally 
accepted  method  of  counting  the  number  of  requests  to  repeat  words  or  sentences,  in  a 
short  interval  of  time,  made  by  talkers  with  average  volume  over  the  circuit  to  be  eval- 
uated. Other  methods  of  evaluation,  such  as  the  "immediate  appreciation"  method, 
have  been  studied  but  have  not  been  adopted  in  the  United  States. 

For  convenience,  each  effective  loss  is  considered  to  have  three  components — volume, 
distortion,  and  sidetone  losses.  The  distortion  and  sidetone  losses  in  the  working  ref- 
erence system  are  considered  to  be  zero  for  reference  purposes,  and  the  effective  loss  of  this 
system  is  thus  numerically  equal  to  the  volume  loss. 

The  individual  effective  losses  which  make  up  the  effective  loss  of  a  complete  circuit 
are: 

1.  Transmitting  loop  loss. 

2.  Receiving  loop  loss. 

3.  Trunk  loss. 

4.  Terminal  junction  loss. 

5.  Central-office  loss. 

6.  Intermediate  junction  loss. 

7.  Loss  due  to  line  noise. 

The  effective  loss  due  to  room  noise  is  not  considered  a  circuit  loss,  although  it  does  affect 
conversations. 

Losses  1  and  2  are  determined  by  comparing  the  effect  on  conversation  of  the  element 
or  complete  loop  to  be  rated  with  the  effect  of  the  corresponding  element  or  complete  loop 
of  the  reference  system,  using  the  other  components  of  the  reference  system  to  complete 


17-80  TELEPHONY 

the  talking  circuit.  Thus,  since  the  effective  loss  of  the  transmitting  loop  of  the  refer- 
ence system  is  7.5  db,  any  transmitting  loop  which  is  substituted  for  the  reference  loop  and 
which  gives  the  same  grade  of  service  (repetition  rate)  also  has  an  effective  loss  of  7.5  db. 
However,  if  the  substituted  loop  gives  the  same  grade  of  service  after  an  increase  of  2  db 
is  made  in  the  variable  reference  trunk  loss,  then  the  substituted  loop  has  an  effective  loss 
2  db  lower  than  the  reference  loop,  or  if  the  reference  trunk  loss  is  adjusted  to  2  db  less  than 
normal  the  substituted  loop  has  an  effective  loss  2  db  higher  than  the  reference  loop. 
Receiving  loops  are  rated  in  the  same  manner,  using  the  reference  receiving  loss  of  1.8  db. 

A  given  trunk  may  be  rated  by  substituting  it  for  the  reference  trunk  and  using  the 
reference  loops  in  the  connection.  The  adjustment  in  the  reference  trunk  loss  deter- 
mines the  rating  of  the  given  trunk  (called  the  effective  connecting  circuit  loss)  with  respect 
to  the  reference  trunk  loss  of  8.7  db,  the  fact  that  the  assigned  effective  loss  of  the  refer- 
ence trunk  is  equal  to  its  attenuation  below  3000  cycles  being  kept  in  mind.  The  ef- 
fective connecting  circuit  loss  thus  obtained  is  not  in  a  useful  form  but  may  be  made  so  by 
dividing  it  into  an  effective  trunk  loss  and  two  effective  terminal  junction  losses  (one  at  each 
end) ,  which  latter  are  considered  equal  for  this  symmetrical  circuit. 

This  division  between  trunk  and  junction  losses  is  necessary  for  the  practical  establish- 
ment of  a  set  of  effective  trunk  loss  curves,  since  for  each  type  of  trunk  the  terminal  junc- 
tion losses  are  different  for  each  different  combination  of  loops,  sets,  and  central  offices  at 
the  trunk  terminals.  The  effective  trunk  loss  consists  of  the  volume  attenuation  of  the 
trunk  and  that  part  of  the  distortion  that  is  proportional  to  trunk  length,  thus  permitting 
the  establishing,  for  each  type  of  trunk,  of  a  value  of  effective  trunk  loss  on  a  per  mile 
basis.  Each  effective  terminal  junction  loss  includes  a  volume  reflection  correction  plus  one- 
half  of  that  part  of  the  distortion  loss  of  the  trunk  that  is  not  included  in  the  effective  trunk 
loss  plus  a  correction  for  the  effect  on  sidetone  of  the  trunk  impedance.  The  effective 
terminal  junction  losses  can  thus  be  considered  as  correcting  factors  which,  when  added  to 
the  sum  of  the  other  losses,  will  give  the  correct  effective  loss  for  a  complete  circuit. 

The  assumption  that  the  effective  connecting  circuit  loss  (trunk  rating)  is  made  up  of  a 
constant  plus  a  loss  proportional  to  length  is  a  good  approximation  for  complete  circuits 
containing  a  single  type  of  trunk,  except  for  effective  trunk  losses  of  about  5  db  or  less 
and  for  coil  loaded  trunks  of  any  effective  loss.  For  the  low-loss  trunks,  accuracy  in  de- 
termining the  trunk  loss  is  not  usually  important.  For  loaded  trunks,  the  curve  of  con- 
necting circuit  loss  versus  length  departs  from  a  straight  line,  because  the  end  sections 
change  with  a  change  in  the  trunk  length.  However,  if  the  connecting  circuit  loss  is 
plotted  only  for  lengths  permitting  half-section  termination,  a  smooth  curve  is  obtained 
which  closely  approximates  a  straight  line  above  5-db  loss,  and  this  curve  may  be  used 
to  determine  the  trunk  loss  per  mile  and  the  terminal  junction  loss  for  this  termination. 
Similar  curves,  approximating  straight  lines,  paralleling  the  one  for  half-section  termina- 
tion, may  be  set  up  for  any  other  desired  end  section  at  each  end.  Such  approximate 
straight  lines  determine  the  effective  terminal  junction  losses  for  the  respective  end  sec- 
tions chosen.  Thus,  the  effective  connecting  circuit  loss  of  a  loaded  trunk,  terminated  at 
half-section  at  one  end  and  at  other  than  half-section  at  the  other  end,  is  considered  as 
made  up  of  the  effective  trunk  loss  per  mile  times  the  total  geographical  length  of  the  trunk 
in  miles  (including  the  end  sections)  plus  the  effective  terminal  junction  loss  for  each  end 
section. 

The  effective  loop  losses  apply  directly  only  with  the  600-ohm  reference  trunk.  The 
effective  trunk  loss  applies  satisfactorily  for  any  combination  of  loop,  set,  and  central 
office,  but  the  effective  terminal  junction  losses  obtained  with  the  reference  loops  do  not 
apply  for  other  combinations  of  loop,  set,  and  central  office.  The  terminal  junction  losses 
for  other  combinations  of  these  elements,  including  other  than  the  reference  trunk,  may 
be  determined  as  required. 

EFFECTIVE  LOOP  LOSSES  include  the  loss  of  three  different  types  of  simplified 
central-office  circuits,  namely,  the  24-volt  repeating  coil  circuit  used  in  the  working  refer- 
ence system,  a  48-volt  repeating  coil  circuit  typical  of  standard  circuits  for  toll  connections, 
and  a  48-volt  step-by-step  circuit  used  in  local  dial  offices. 

In  practice,  actual  central-office  circuits  have  equipment  and  wiring,  additional  to  the 
simplified  circuits,  the  loss  of  which  is  dependent  upon  the  actual  loop  and  trunk  condi- 
tions. However,  a  single  value  of  loss  for  the  additional  equipment  and  wiring  for  each 
type  of  office  and  connection  is  usually  sufficient  if  it  is  determined  under  typical  limiting 
loop  conditions.  The  effective  local  offices  losses  are  therefore  determined  between  the 
working  reference  loops  and  trunk  for  the  more  commonly  used  central-office  connecting 
circuits. 

For  trunks  composed  of  different  types  of  facilities,  such  as  19-  and  22-gage  cable, 
intermediate  junction  losses  occur  at  each  junction  of  the  dissimilar  types  of  facilities  be- 
cause of  reflections  of  energy  at  these  points.  Such  losses  have  been  determined  for  var- 


SERVICE  REQUIREMENTS — EXCHANGE 


17-81 


ious  combinations  of  facilities  used  in  the  Bell  System  and  must  be  added  as  part  of  the 
overall  trunk  loss. 

Effective  losses  due  to  line  noise  (Table  3  of  this  section)  should  be  added  to  the  other 
effective  losses  in  a  given  loop  when  the  electrical  noise  at  the  receiver  terminals  of  the 
loop  is  greater  than  the  reference  noise  (100  units)  assumed  in  the  reference  loops.  Less 
noise  than  reference  noise  in  the  receiver  is  considered  equivalent  to  reference  noise. 

In  addition  to  the  above-mentioned  losses,  if  the  room  noise  for  a  given  loop  is  more  or 
less  than  the  room  noise  (50  db  RAP)  assumed  for  the  reference  loop,  effective  room  noise 
losses  corresponding  to  the  difference  between  the  actual  and  assumed  room  noise  must 
also  be  added  to  the  overall  given  loop  loss.  Figure  12  shows  the  effective  loss  due  to 
room  noise  versus  room  noise  of  different  intensities  for  two  different  sidetone  conditions. 


Il 


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Sg  I 
il  § 


3    15 


-5 


A 

7 

A 

f 

S.HIGH  SIDETONE, 
NO.  46  INDUCTION  CO! 
STANDARD  CONNEC- 
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w 

m 

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M 

^A,  LOW  SIDETONE, 
ANT1-SIDETONE 
INDUCTION   COILS 

A 

7 

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f 

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^^ 

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-20      -15      -10      -5          O         5         10        15        20       25        3O 
ROOM  NOISE  IN  DECIBELS  ABOVE  TYPICAL  VALUE 


35 


FIG.  12.    Effective  Receiving  Loss  Due  to  Room  Noise,  versus  Room  Noise  of  Different  Intensities 

(Courtesy  Bell  System) 

The  values  shown  are  approximate,  since  it  is  difficult  to  determine  the  actual  losses  due 
to  room  noise  because  of  a  number  of  variable  factors,  some  of  an  intangible  nature. 

The  Bell  Telephone  Laboratories  have  made  field  and  laboratory  investigations  com- 
paring the  ease  of  carrying  on  a  telephone  conversation  over  different  circuits  in  actual 
service,  and  also  studying  the  physical  characteristics  of  circuits  and  instruments,  as  well 
as  their  ability  to  transmit  speech  sounds,  using  syllabic  articulation  tests  under  a  large 
number  of  variable  conditions  encountered  in  service.  From  these  investigations  a 
number  of  effective  transmission  loss  curves  showing  effective  loop  losses  in  decibel  versus 
loop  length  in  thousands  of  feet  have  been  prepared  for  the  commonly  used  types  of  cen- 
tral offices,  loop  facilities  (cable  or  open  wire) ,  and  telephone  sets,  based  on  the  working 
reference  system. 

Figure  13  shows  separate  effective  transmitting  and  receiving  loss  curves  (provisional) 
for  both  24-volt  exchange  grade  and  48-volt  toll  grade  battery  supply  and  also  the  average 
(T  +  R)/2  curves  for  both  grades  of  battery  supply.  These  particular  curves  apply  for 
W.E.  Co.  type  1  or  10  Manual  or  Panel  Dial  Offices,  non-loaded  22-BSA  gage  cable  loops, 
and  the  latest-type  W.E.  Co.  Antisidetone  (AST)  subscriber  sets  with  FIA  (handset)  or 
635-706A  (deskstand)  type  instruments.  These  curves  are  typical  of  other  sets  of  sub- 


17-82 


TELEPHONY 


scriber  loop  loss  curves  for  different  types  of  central  offices,  loop  facilities,  and  sets,  ex- 
cept that  the  loss  values  are  different  for  the  different  conditionso 

In  addition  to  the  non-loaded  subscriber  loop  loss  curves,  effective  loss  curves  are  re- 
quired showing,  for  different  lengths  and  conditions,  the  losses  of  loaded  trunks  and  loops, 
subscriber  loop  losses  having  loops  composed  of  two  or  more  different  types  of  facilities, 
current  supply  losses  versus  transmitter  current  and  versus  loop  resistance,  transmitting 
and  receiving  losses  due  to  sidetone,  terminal  junction  losses,  and  many  similar  curves 
useful  to  the  engineer. 


LOOP  LENGTH  IN  THOUSANDS  OF  FEET 
,00  0  0  &  0  0  C 

R«  RECEIVING  LOSS 
..   T=  TRANSMITTING  LOSS 
SUBSCRIPTS: 

Es34V  EXCHANGE  GRADE 
BATTERY  SUPPLY. 
-     Ts  48V  TOLL  GRADE 
BATTERY  SUPPLY. 
600-OHM  REFERENCE  TRUNK 

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2-10-6-6-4-2            0            24             6            &            K>           12          14            16           M 

EFFECTIVE  LOOP  LOSS  IN   DECIBELS 

NO.  1  OR  NO.  10  MANUAL  OR  PANEL  DIAL  OFFICE  F1A- AST\ 

32-BSA   CABLE  LOOP  635-70«A-  AST/ 

FIG.  13.    Effective  Transmission  Losses  in  Common-battery  Subloops  (Courtesy  Bell  System) 

SWITCHBOARD  OPERATOR  EFFECTIVE  TRANSMISSION  PERFORMANCE  is 

also  rated  by  means  of  the  working  reference  system,  except  that  the  reference  trunk  im- 
pedance is  900  ohms  instead  of  the  600  ohms  employed  for  subscriber  loop  ratings.  In 
making  comparisons,  the  operator  circuit  is  connected  to  the  reference  system  in  place  of 
one  of  the  subscriber  loops.  Since  no  loop  is  involved  for  this  connection,  single  values  of 
transmitting  and  receiving  losses  are  adequate  for  rating  each  combination  of  operator's 
telephone  set  circuit  and  instruments. 

The  typical  values  of  line  and  room  noise  specified  for  the  operator  terminal  are  45  db 
RAP  for  line  noise  in  the  listening  ear  and  65  db  RAP  for  room  noise.  The  reference 
operator  receiver  (W.E.  Co.  No.  528)  is  more  efficient  than  the  reference  subscriber  re- 
ceiver (W.E.  Co.  No.  144) ,  and  the  reference  room  noise  for  operators  is  higher  than  that 
for  subscribers.  Incoming  room  noise  from  the  distant  operator's  set  also  adds  to  the 
overall  effective  loss  in  the  receiving  operator's  circuit.  The  latest  design  in  operator 
transmitters  and  receivers  provides  improvements  in  operator  transmission  which  will 
permit  overall  operator  to  operator  standards  in  the  approximate  range  of  10  to  12  db, 
which  is  comparable  with  present  local  subscriber  to  subscriber  plant  design. 


16.  PLANT  DESIGN— TOLL 

TOLL  FACILITIES  have  been  developed,  since  the  earliest  open-wire  (iron)  type,  to 
include  the  following:  (1)  open  wire,  (2)  cable,  (3)  carrier,  and  (4)  radio. 

TOLL  OPEN-WIRE  FACILITIES  consist  principally  of  hard-drawn  copper,  copper 
steel,  and  high-tensile-strength  steel  bare  wire. 

The  three  gages  of  hard-drawn  copper  line  wire  employed  most  commonly  in  Bell  Sys- 
tem plant  are  165,  128,  and  104  (mil  diameter)  wire,  the  electrical  characteristics  of  which 
are  shown  for  various  positions  and  arrangements  on  the  pole  line  in  Fig.  14.  For  wet 
weather  the  value  of  g  increases  and  other  factors  change.  Several  other  gages  of  copper 
line  wire  are  in  use  in  this  and  other  countries,  such  as  102  mil  diameter,  but  the  three 
mentioned  above  are  representative  of  this  type  of  wire  usage.  Also,  a  number  of  basic 
wire  gages  are  in  use,  so  that  the  mil  diameter  of  a  given  gage  of  wire,  such  as  No.  8,  may 
be  different  under  the  different  basic  gages.  Figures  15,  16,  and  17  show,  respectively, 
the  attenuation  frequency  characteristics  of  different  gages  of  hard-drawn  copper  physi- 


PLANT  DESIGN TOLL 


17-83 


5 


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17-84 


TELEPHONY 


cal  (side  in  Fig.  15)  circuits  over  the  voice,  type  C,  and  type  J  carrier  frequency  ranges, 
for  both  wet  and  dry  weather  conditions.  The  wire  spacing  and  types  of  insulators 
involved  are  12-in.  and  double-petticoat  (DP)  for  Fig.  15,  and  8-in.  and  CS  glass  for 
Figs.  16  and  17.  In  the  frequency  range  of  20  to  150  kc,  the  attenuation  factor  in- 
creases rapidly  with  frequency  where  the  wires  are  covered  with  snow  or  ice.  For  example, 


0         OS        1.0       1.5       2.0       2J       3.O      a5       4.O 
FREQUENCY  IN   KILOCYCLES  PER   SECOND 

FIG.  15.      Attenuation-frequency  Characteristics  of  Open-wire  Side  Circuits  over  the  Voice  Range 

(Courtesy  Bell  System) 

with  about  1/3  in.  total  diameter  of  melting  glaze  on  a  type  J,  8  in.,  CS  insulated,  165- 
gage  carrier  pair,  the  attenuation  increases  approximately  from  0.13  db  per  mile  at  20 
kc  to  0.9  db  per  mile  at  150  kc.  Variations  of  attenuation  with  temperature  due  to  re- 
sistance change  in  open  wire  are  about  1  per  cent  per  4  i/2  deg  fahr  change  from  68  deg 
fahr. 

In  recent  years,  copper  steel  wire  has  been  used  extensively  for  telephone  circuits,  com- 
bining strength  with  relatively  low  transmission  losses  and  d-c  resistance  values.  This 
wire  is  manufactured  with  30  and  40  per  cent  conductivity,  which  is  the  conductivity 
ratio  (in  percentage)  of  the  wire  to  that  of  the  annealed  copper  standard  of  like  diameter. 


O.22 
0.20 
0.1S 
O.I  6 
0.14 
0.12 
O.IO 
0.08 
0.06 
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y, 

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s 

>         5         10        15       20       25       30       35       40      45      5< 
FREQUENCY  IN   KILOCYCLES  PER   SECOND 

FIG.  16.     Attenuation-frequency  Characteristics  of  Open-wire  Physical  Circuits  over  the  Type  C 
Carrier  Range  (Courtesy  Bell  System) 

A  40  per  cent  conductivity  copper  steel  wire,  which  is  the  more  commonly  used  type,  has  a 
steel  core  with  a  welded  copper  casing  having  a  radial  thickness  of  20  per  cent  of  the  total 
radius  of  the  wire. 

The  tensile  strength  and  attenuation  of  copper  steel  pairs  are  about  2  to  2  1/2  times  that 
of  hard-drawn  copper  of  the  same  size.     Owing  to  higher  attenuation,  telephone  repeaters 


PLANT  DESIGN — TOLL 


17-85 


2  0.32 


<  0.16 


WET  WEATHER 

DRY  WEATHER 


40  60  SO  1OO          120          140 

FREQUENCY  IN  KlUOCYCLES  PER  SECOND 


FIG.  17.     Attenuation-frequency  Characteristics  of  Open-wire  Physical  Circuits  over  the  Type  J 
Carrier  Range  (Courtesy  Bell  System) 


require  closer  spacing  with  copper  steel  than  with  hard-drawn  copper  circuits, 
acteristics  of  copper  steel  pairs  are  given  in  Table  5. 


The  char- 


Table  5.    Characteristics  of  Copper  Steel  Pairs 
(Estimated  for  68  deg  Fahr — 40  Per  Cent  Conductivity — 53  Pairs  CS  Insulators  per  Mile) 


Size  of 

Fre- 

Resist- 

Inductance, 
mh  per  pair 
Tnile 

Characteristic 
Impedance,  Dry,  ohms 

Attenuation, 
db  per  mile, 
dry 

Attenuation, 
db  per  mile, 
wet 

Wire, 

mil 
diameter 

kc 

ohms 

Pin  Spacing 

Pin  Spacing 

Phi  Spacing 

Pin  Spacing 

8  in. 

12  in. 

8in, 

12  in. 

Sin. 

12  in. 

8  in. 

12  in. 

165 

0.2 

9.9 

3.125 

3.385 

761-J516 

793-J542 

0.057 

0.054 

0.063 

0.060 

1.0 

10.3 

3.060 

3.320 

572-/I43 

615-jl43 

0.078 

0.073 

0.084 

0.078 

3.0 

10.8 

3.027 

3.287 

558  -j  51 

597  -j  51 

0.085 

0.079 

0.092 

0.086 

10.0 

11.4 

2.995 

3.255 

552  -j  16 

592  -j  16 

0.092 

0.085 

0.104 

0.098 

30.0 

12.6 

2.986 

3.246 

551  -j    6 

591  -  j   6 

0.103 

0.095 

0.127 

0.120 

140.0 

24.7 

2.974 

3,234 

550  -j    2.4 

590  -j   2.4 

0.207 

0.192 

0.282 

0.273 

128 

0.2 

16.6 

3.285 

3,545 

943-J736 

991  -  ;759 

0.077 

0.073 

0.084 

0.081 

1.0 

17.3 

3.250 

3.510 

635-;230 

674-J232 

0.120 

0.112 

0.126 

0.119 

3.0 

18.0 

3.206 

3.466 

593  _  j  86 

634  -j  86 

0.134 

0.124 

0.142 

0.132 

10.0 

18.7 

3.167 

3.427 

583  -j  27 

625  -  j  27 

0.142 

0.131 

0.155 

0.145 

30.0 

19.7 

3.148 

3.408 

580  -j    9.4 

623  -j    9.4 

0.152 

0.139 

0.177 

0.168 

140.0 

30.5 

3.139 

3.399 

580  -j    3 

622  -j    3 

0.241 

0.224 

0.321 

0.310 

104 

0.2 

25.0 

3.430 

3.690 

1139  -;958 

1189  -J988 

0.096 

0.091 

0.105 

0.101 

1.0 

25.7 

3.410 

3.670 

691  -  J324 

736-J328 

0.162 

0.152 

0.169 

0.160 

3.0 

26.6 

3.357 

3.617 

621  -  j\24 

666-J124 

0.188 

0.174 

0.1% 

0.183 

10.0 

27.7 

3.313 

3.573 

606  -j  40 

651  -j  40 

0.201 

0.186 

0.215 

0.200 

30.0 

28.7 

3.287 

3.547 

602  -j  14 

647  -  j  14 

0.211 

0.195 

0.238 

0.224 

140.0 

37.6 

3.277 

3.537 

602  -j    3.8 

647-;    3.8 

0.284 

0.264 

0.367 

0.353 

Notes: 

1.  Resistance  (d-c)  is  0.1  ohm  less  than  resistance  at  0.2  kc  for  all  three  gages. 

2.  Leakage  conductance  and  capacities  of  copper  steel  pairs  are  comparable  to  those  of  same  size  hard-drawn  copper 
pairs  (Fig.  14). 

3.  For  DP  insulators  the  attenuation  change  from  dry  to  wet  weather  conditions  is  about  twice  that  for  CS  insulators. 

4.  The  above  estimated  values  will  vary  somewhat  from  actual  measurements,  owing  to  the  effects  of  transpositions, 
spacing,  and  other  small  irregularities,  which  cannot  be  calculated.    For  12-In.  pairs,  assume  on  the  average  about 
10  per  cent  lower  high-frequency  impedance  and  about  10  per  cent  higher  high-frequency  attenuation  values.    The 
deviations  should  be  less  for  the  closer  spaced  pairs  with  point-type  transpositions.    (Courtesy  Bell  System.) 


17-86 


TELEPHONY 


HIGH-STRENGTH  STEEL  WIRE  has  practically  replaced  the  various  grades  of  iron 
(E.B.B.  and  B.B.)  and  mild  steel  wire  for  telephone  purposes  because  of  its  greater  strength 
and  coordination  with  other  wire  services.  The  high-strength  steel  wire  now  in  production 
by  several  manufacturers  is  used  for  only  the  very  short  toll  circuits  in  some  parts  of  the 
country,  on  account  of  its  high  attenuation  and  inherent  noise  characteristics.  Its  prin- 
cipal usage  is  in  exchange  plant.  Steel  wire  is  now  generally  zinc  coated  (galvanized) 
electrolytically,  which  insures  a  more  uniform  coating  than  the  "hot  dip"  process.  The 
usual  weights  of  zinc  coating  vary  from  0.8  to  2.4  oz  per  sq  ft,  depending  upon  the  custom- 
er's requirements.  The  characteristics  are  given  in  more  detail  in  article  17,  Plant  De- 
sign— Exchange. 

CABLE  FACILITIES  for  toll  purposes  comprise  a  number  of  types  of  loaded  and  non- 
loaded  cable  pairs.  Toll  entrance  cables  are  employed  for  extending  open-wire  toll  lines 

into  toll  or  toll  terminal 

Table  6.     Load-coil  Spacings  offices  or  stations  for  dis- 

tances usually  limited  to  a 
few  miles.  Toll  cables  are 
employed  as  permanent 
backbone  toll  plant,  inter- 
connecting principal  cities 
and  important  intermedi- 
ate switching  and  equip- 
ment centers.  Toll  and 
toll  entrance  cables  may 
be  of  the  aerial  or  under- 
ground type  or  combina- 
tions of  both,  but  the 
tendency  is  to  place  them 
underground  for  greater 
reliability  of  service. 

Cable    facilities,    being 
*For  side  and  phantom  cable  capacitances  of  0.062  and  0.102  /*£    necessarily  of  small-gage, 


Code 
Designation 

Nominal 
Spacing, 
feet 

General  Application 

A 

700* 

Cables  serving  open-wire  carrier. 

B 

3000* 

Cables  serving  open-wire  carrier.    Toll  and 

exchange  cables. 

C 

929* 

Cables  serving  open-  wire  carrier. 

D 

4500* 

Exchange  cables. 

E 

5575* 

Toll   entrance   cables.      Replaced   by    H, 

except  when  used  with  C  spacing. 

F 

2787* 

Cables  serving  open-wire  carrier  (phantom 

circuits). 

H 

6000* 

Toll,  toll  entrance,  and  exchange  cables. 

J 

600  t 

Cables  serving  J  open-wire  carrier. 

X 

680* 

Equivalent  capacity  in  carrier  office  cables. 

Y 

2130  * 

Equivalent  capacity  in  carrier  office  cables. 

per  mile,  respectively. 

t  For  physical  pair  capacitance  of  0.025  /if  per  mile. 


soft-drawn,  insulated  cop- 
per conductors,  in  order  to 
provide  economical  sizes 
of  conductor  complements,  have  relatively  high  mutual  capacitances,  resistances,  and 
greatly  increased  attenuation  per  mile  over  the  larger-gage,  wider-spaced,  open-wire  con- 
ductors. Figure  18  shows  the  characteristics  of  standard  types  of  paper-insulated  cable 
telephone  circuits  at  1000  cycles  per  second,  as  used  in  the  Bell  System.  In  column  2  of 
Fig.  IS,  the  abbreviations  N.L.S.  and  N.L.P.  refer  to  non-loaded  side  and  non-loaded 
phantom  respectively,  while  the  remaining  abbreviations  show  the  load  coil  spacing  by 
letter  (H-6000  feet  and  B-3000  ft),  the  inductance  in  millihenries  by  the  figure,  and  whether 
the  circuit  is  a  side  or  phantom  by  the  letters  S  and  P  respectively.  For  a  designation 
such  as  H-44-25,  the  first 

and  second  figures  refer  to  Table  7.     Toll-entrance  Cable  Loading 

the  side  and  phantom  coil 
inductances,  respectively. 

The  type  of  loading  at 
present  employed  in  toll 
cables  for  four-wire  voice 
frequency  message  cable 
circuits  is  usually  the  19- 
gage  H-44-25  or  H-88-50 
type,  since  less  variation 
in  attenuation  over  the 
voice  range  and  higher 
cutoff  frequencies  are  ob- 
tained with  these  loadings 
(Fig.  18)  than  are  possible 
with  the  older,  higher- 
inductance-type  loading.  Repeaters  generally  are  spaced  at  40-50  mile  intervals.  For 
two-wire  voice-frequency  message  cable  circuits,  19-gage  H-88-50  or  B-88-50  loading  is 
usually  employed.  The  variation  in  attenuation  over  the  voice  range  is  small,  the  cutoff 
frequency  is  amply  high,  and  telephone  repeater  spacings  of  about  40-50  miles  generally 
are  used. 

For  program  transmission  (article  18)  16-gage  B-22  loaded  cable  pairs  were  specially 


Carrier 
Frequency 
Range,  kc 

Type  of  Loading 

0-   10 
0-  30 

0-  30 
0-145 

0-145 
0-145 
0-  30 
0-  10 

B  H-15-15  (B  for  side,  H  for  phantom) 
C  E-4.1-12,8  (C  for  side,  E  for  phantom  and  1  65  open- 
wire  circuits,  1  2  in.) 
C  E-4.8-1  2.8  (C  for  side,  E  for  phantom  and  1  28  or  1  04 
open-wire  circuits,  1  2  in.) 
J-0.72         1  65  open-wire,  8  in.  i  Used    with    disk-insu- 
128  open-wire,  8  in.  I      lated  pairs  in 
J-0.85         1  28  open-wire,  8  in.  |      shielded,    spiral-four 
J-0.94         104  open-wire,  8  in.  J      quadded  cable. 
X-2.7         Office  cable  loading. 
Y-9             Office  cable  loading. 

PLANT  DESIGN TOLL 


17-87 


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17-88 


TELEPHONY 


developed  to  give  a  high  cutoff  frequency.  These  facilities  are  capable  of  transmitting  a 
frequency  band  up  to  about  8000  cycles  without  serious  distortion. 

Figure  19  shows  the  attenuation  frequency  characteristics  of  various  types  of  cable 
circuits  loaded  for  voice  frequency  and  program  service. 

Table  6  gives  the  code  designations  of  load  coil  spacings  as  devised  for  Bell  System  use. 


x** 

19  GAUGE  H-I74-106 
CIRCUITS 

/ 

S^ 

.—       •" 

^ 

SID 

E  X* 

/ 

(-" 

,x--- 

PHAf 

slTOM 

/- 

•*    •— 

_-  —  •  - 

^*.** 
>HAh 

.x 

mDM 

1 

19  GAUGE  H-44-25 
CIRCUITS 

j        0        0.5       1.0       1.5      2.O     2.5      3.0      3.5    0       0.5       1.0       1.5      2.0      2.5      3.0      3.5      4.0     4.5      5.0 


SIDE 


19  GAUGE  H-88-50 
CIRCUITS 


O        0.5       1.0       4.5      2.0      2.5      3.0      3.5     0       0.5       1.0       1.5      2.0      2.5     3.0      3.5      4.0      4.5     5.0 


/ 

SID 

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E/ 

/  / 

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—  —  • 

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..  ^TPHAh 

JTOM 

16  GAUGE  B-88-5O 
CIRCUITS 

1            i            1 

5          6          70         1          23          4          5 
FREQUENCY  IN  KILOCYCLES   PER  SECOND 


PIG.  19.    Attenuation-frequency  Characteristics  of  Various  Types  of  Loaded  Cable  Circuits  (Courtesy 

Bell  System) 

The  type  of  loading  generally  used  in  toll  entrance  cables  is  of  the  H-31-18  type  for  voice- 
frequency  circuits  and  of  the  types  shown  in  Table  7  for  carrier  circuits. 

Non-loaded  cable  facilities  are  employed  in  both  toll  entrance  and  toll  cables.  Such  toll 
entrance  facilities  may  be  used  without  appreciable  transmission  penalty  up  to  about 
3000  ft  in  extending  open-wire  circuits  into  offices  or  intermediate  toll  equipment  points. 
For  K  carrier  systems,  which  are  operated  through  toll  cables,  only  non-loaded  pairs  may 
be  used,  since  loading  is  not  available  for  the  K  frequency  range.  For  J  carrier  systems  a 
specia%  designed  low-capacity,  16-gage,  spiral-four  loaded  or  non-loaded  cable  may  be 
used  for  entrances. 


PLANT  DESIGN — TOLL 


17-89 


Cable  faculties  used  for  toll  purposes  have,  in  general,  been  composed  of  10-,  13-,  16-, 
and  19-gage  cable  conductors,  paired  and  quadded  for  physical  and  phantom  circuit  oper- 
ation. Some  non-quadded  cables  for  physical  circuits  only  have  been  used.  At  present 
19  gage  is  largely  employed,  but  in  a  few  cases  16  gage  may  be  added.  The  newer  toll 
cables  have  the  short  pair  twist,  high  dielectric  strength,  core  to  sheath,  and  nominal 
mutual  capacities  of  0.062  and  0.096  /if  per  mile  for  side  and  phantom  respectively.  The 
standard  sizes  of  toll  and  toll  entrance  cable  cover  a  wide  range  of  standard  numbers  of 
quads  and  pairs,  and  several  different  gages  may  be  placed  in  the  same  lead  cable  sheath, 
including  pairs  for  exchange  use,  as  required  to  meet  service  needs. 

Toll  entrance  cables  for  carrier,  non-phantomed,  open-wire  lines  have  the  long  pair 
twist  and  nominal  capacities  of  0.062  and  0.102  juf  per  mile  for  the  side  and  phantom, 
respectively. 

Variations  in  net  losses  in  cable  facilities  due  to  temperature  changes  are  shown  in  Fig. 
20. 


Loading 

Loss  at  55  Deg  Fahr,  decibels  per  mile 

22-gage 

19-gage 

I6-gage 

13-  or  14- 
gage  * 

10-gage 

B-22-N 
B-44-N 

B-88-50-S 
B-88-50-P 

H-22-N 

H-44-25-S 
H-44-25-P 

H-88-50-S 
H-88-50-P 

H-172-63-S) 
H-174-63-SJ 

H-172-63-P) 
H-174-63-PJ 

H-174-I06-S 
H-174-106-P 

0.45 
.34 

0.236 

0.157 

a.  so 

.42 

.28 
.23 

.62 

.47 
.39 

.35 

.30 

.27 

.28 

.28 
.22 

.16 
.14 

.32 

.25 
.21 

.19 
.16 

.16 

.16 

.16 
.13 

.92 
.77 

.66 
.56 

.49 

.51 

.50 
.40 

.14 
.115 

.12* 

0.08 
.07 

.10 
.084 

*  Value  marked  with  *  applies  to  14-gage. 


Loading 

±  Variations  from  Loss  at  55  Deg  Fahr,  decibels  per  mile 

22-gage 

19-gage 

I6-gage 

Aerial 
Cable  * 

U.G. 

Cables  t 

Aerial 
Cable  * 

U.G. 

Cable  f 

Aerial 
Cable  * 

U.G. 

Cable  f 

B-22-N 
B-44-N 

B-88-50-S 

0.052 
.040 

.031 

0.017 
.013 

.011 

0.028 

0.009 

0.060 

0.020 

.018 

.006 

B-88-50-P 
H-22-N 

H-44-25-S 
H-44-25-P 

.050 

.017 

.026 
.071 

.055 
.046 

.009 
.024 

.018 
.015 

.015 
.037 

.029 
.024 

.005 
.012 

.010 
.008 

.110 
.092 

.037 
.031 

H-88-50-S 
H-88-50-P 

.079 
.067 

.026 
.022 

.041 
.035 

.014 
.012 

.022 
.019 

.007 
.006 

H-172-63-S1 
H-174-63-SJ 

.059 

.020 

.031 

.010 

.018 

.006 

H-172-63-P) 
H-174-63-PJ 

.061 

.020 

.032 

.011 

.018 

.006 

H-174-106-S 
H-174-106-P 

.060 
.048 

.020 
.016 

.032 
.025 

,011 
.008 

.017 
.013 

.006 
.004 

*  Temperature  range,  ±54  deg  fahr;  resistance  variation,  ±12  per  cent 
f  Temperature  range,  ±18  deg  fahr;  resistance  variation,  ±4  per  cent 

FIG  20      Net  Loss  Variations  with  Temperature  for  Different  Gages  and  Loadings  of  Cable  Circuits 

(Courtesy  Bell  System) 


17-90 


TELEPHONY 


The  cables  may  have  a  plain  lead  sheath  covering  with  no  outer  protection,  or,  if  pro- 
tection is  required  from  gophers,  digging,  or  other  extraneous  disturbances,  the  sheath 
may  be  covered  with  jute,  gopher  and  jute  protection,  corrosion  protection,  or  tape  armor 
for  either  aerial  or  buried  construction.  Submarine  cables  have  single  or  double  armored 
protection,  composed  of  heavy  steel  wires  laid  around  the  sheath  with  jute  and  compound 
in  one  or  two  layers.  Metallic  shields  are  also  used,  as  required,  around  the  cable  core, 
individual  groups  or  quads  of  cable  conductors,  to  protect  circuits  from  electrical  dis- 
turbances. 

CARRIER  FACILITIES  may  be  provided  in  either  open  wire  or  cable.  For  any  given 
carrier  system  between  two  points,  the  facilities  selected  must  be  suitable  for  that  system. 
Low-frequency  carrier  systems  of  the  G  and  H  types  generally  employ  12-in.-spaced,  open- 
wire  facilities,  having  voice-frequency  characteristics,  except  that,  as  the  number  of  these 
systems  increase  on  a  given  pole  line,  a  carrier  transposition  scheme  may  be  required  to 
limit  crosstalk,  since  the  usual  voice-frequency  schemes  are  not  designed  for  the  higher 
frequencies.  The  C  carrier  system  generally  employs  12-  or  8-in.-spaced,  open- wire 
facilities,  transposed  for  frequencies  up  to  30  kc,  although  the  physical  pairs  or  phantom 
groups  may  be  transposed  one  at  a  time,  according  to  the  selected  transposition  scheme, 
as  required  to  provide  for  the  carrier  systems  on  the  given  pole  line.  The  J  carrier  system, 
being  in  the  high-frequency  group,  requires  8-  or  6-in.-spaced,  open-wire  facilities  specially 
transposed  to  limit  crosstalk  at  frequencies  up  to  140  kc. 

With  the  G,  H,  and  C  systems,  DP  insulators  and  transposition  drop  brackets  are  usu- 
ally satisfactory  for  wet  or  dry  weather,  but  for  the  J  system  it  is  necessary  to  provide 
CS  insulators  and  point-type  transpositions,  where  the  pole  line  will  initially  or  ultimately 
have  a  full  complement  of  carrier  systems.  Wire  sag  must  be  limited  to  small  deviations 
from  normal  and  all  other  physical  irregularities  must  be  controlled  where  carrier  systems 
are  operated.  These  limitations  assume  greater  importance  as  the  system  frequency  in- 
creases. 

The  K  carrier  system,  operating  through  cable  only,  requires  suitable  conductors.  Non- 
loaded  pairs  of  0.062-juf  capacity  per  mile  and  of  19  gage,  selected  as  the  most  economical 
gage,  must  be  properly  balanced  in  the  cable  mutually  and  against  other  facilities  and 
segregated  (using  separate  cables  or  layer  shields)  for  the  two  directions  of  transmission. 
Figure  21  shows  the  attenuation-frequency  characteristics  of  19-gage  non-loaded  cable 
circuits  over  the  K  carrier  range  at  different  temperatures. 


4.2 
4.0 
3.8 
3.6 

i« 

uj  3.2 

a. 

3   3-0 

UJ 

00 

2.6 
2.4 


gx 


8        12       16       20      24      28      32      36      40      44     48      52      56      6O 
FREQUENCY  IN   KILOCYCLES  PER   SECOND 

FIG.  21.    Attenuation-frequency  Characteristics  of  Non-loaded  19-gage  Cable  Circuits  over  the  Type  K 
Carrier  Range  at  Different  Temperatures  (Deg  Fahr)  (Courtesy  Bell  System) 

COAXIAL  CABLE  is  required  for  L  carrier  system  operation,  on  account  of  the  high- 
frequency  range  of  this  system  (up  to  possibly  7  Me  or  more).  This  type  of  cable,  first 
installed  commercially  between  Minneapolis,  Minn.,  and  Stevens  Point,  Wis.,  in  1941,  is 
built  up  of  multiples  of  two  coaxial  tubes  or  units,  plus  ordinary  paper-insulated  con- 
ductors as  needed  to  take  care  of  requirements  for  short-haul  message  circuits  and  signal- 
ing and  alarm  trunks. 


PLANT  DESIGN — TOLL 


17-91 


The  Stevens  Point-Minneapolis  cable 
contained  four  coaxial  units,  with  two 
22-gage  pairs  in  the  center,  a  19-  and 
22-gage  pair  in  each  of  the  four  outer 
interstices  between  the  units,  and  eight- 
een 19-gage  quads  surrounding  this  as- 
sembly. Since  that  installation  many 
changes  and  improvements  have  been 
made  in  the  design  and  construction  of 
coaxial  cable.  One  of  the  latest  designs 
provides  for  8  coaxial  units  and  about 
78  quads  of  19-gage  copper  equivalent, 
built  in  with,  and  surrounding  the  units 
to  form  a  full-size  lead-sheathed  cable, 
The  quads  may  be  used  for  message  and 
program  service,  order  wires,  alarms,  and 
miscellaneous  applications,  and  the  co- 
axial units  for  message,  television,  and 
possibly  other  services. 

The  present  coaxial  unit  consists  of  a 


X 


60 


200         400  600    1000       2OOO       4OOO 
FREQUENCY  JN  KJLOCYCLES  PER  SECOND 


FIG.  22.    Attenuation-frequency  Characteristics  of  an 

0.375-in.  Longitudinal  Seam   Coasiai   Cable  Circuit 

(Courtesy  Bell  System) 


Z76! 


single  12-mil  copper  tape  (outer  conductor)  formed  into  a  tubular  conductor,  which  is,  in 
turn,  tightly  wrapped  with  two  6-mil  steel  tapes,  the  outer  of  which  covers  the  gaps  be- 

tween the  turns  of  the  inner  steel  tape. 
Each  edge  of  the  copper  tape  has  small 
serrations  which  engage  the  opposite 
edge  of  the  copper  tape,  and  the  two 
edges  thus  interlock  to  form  a  tube. 
This  design  is  known  as  "longitudinal 
seam"  coaxial.  A  10-gage  (100.4-mil) 
copper  wire  (inner  conductor)  is  sup- 
ported in  the  center  of  this  tube  by 
polyethylene  disks  (0.085  in.  thick), 
spaced  about  1  in.  apart. 

The  inside  diameter  of  the  present 
standard  coaxial  tube  is  0.375  in.  (an 
original  design  employed  a  13-gage  cen- 
tral conductor  and  an  outer  conductor 
having  an  inside  diameter  of  0.27-in.). 
Attenuation  and  impedance  character- 
istics are  shown  in  Pigs.  22  and  23.  The 
new  type  permits  lengthening  auxiliary 
repeater  spacings  from  about  5.4  miles 
(for  the  0.27-in.  tube)  to  about  7,9  miles 


O-2 

5 

X-3 


2OO          -4OO  6OO     KXXJ        20OO        4OOO 
FREQUENCY  D4   KILOCYCLES  PER  SECOND 


FIG.  23.     Impedance  Characteristics  of  an  0.375-in. 

Longitudinal  Seam  Coaxial  Cable  Circuit  (Courtesy 

Bell  System) 


(for  the  0.375-in.  tube),  with  480  circuits  provided  in  each  case.     Also  the  maximum  spac- 
ing between  main  repeater  stations  is  increased  from  about  90  to  more  than  150  miles. 
The  larger  coaxial  will  transmit  from  1.5  to  2  times  as  wide 
a  frequency  band  as  the  smaller  coaxial,  assuming  the  same 
repeater  spacings,  thus  permitting  the  handling  of  high 
definition  or  color  television  if  the  demand  for  it  arises. 
Figure  24  shows  construction  of  an  eight-unit  coaxial  with 
a  complement  of  wire  quads. 

Lightning  protection  for  the  complete  cable  is  provided 
by  placing  a  10-mil,  corrugated,  copper  jacket  over  the 
lead  sheath  of  the  cable,  which  is  first  wrapped  in  thermo- 
plastic material  and  a  layer  of  tough  cloth.  The  copper 
jacket  is  then  covered  with  cloth  tape. 

Each  regular  coaxial  unit  is  paired  with  an  identical 
alternate  coaxial  unit,  which  is  automatically  switched 
into  service,  replacing  the  regular  unit,  in  case  of  fault 
occurrence. 

PHANTOM  CIRCUIT  operation  is  common  in  both 
open  wire  and  cable.  Two  open-wire  physical  circuits  or 
two  cable  pairs,  which  are  twisted  together  to  form  a  quad 
(four  wires) ,  can  be  equipped  at  their  terminals  to  provide 
a  third  circuit,  called  a  phantom,  as  shown  in  Fig.  25. 


An    8-unit    Coaxial 


2. 

§tment 

tesy  Bell  System) 


17-92 


TELEPHONY 


The  side-circuit  currents  circulate  over  the  side  circuits  as  shown  by  the  dashed  arrows, 
and  the  phantom-circuit  currents  circulate  over  the  phantom  circuit  as  shown  by  the 
solid  arrows.  The  line  sides  of  the  phantom  repeating  coils  are  closely  balanced  with 
respect  to  the  phantom  tap  so  that  the  phantom-circuit  current,  dividing  about  equally 


Office  A                                               Una                                          Office  B 

Repeating 
Coil 

Wire  No.  1 

Repeating 

Coil 

>,      g 
Side  Circuit  1 
,<-  *      «= 

if 

^._.      ^         Wire  No.  2 

41 

!|  Side  Circuit 

—  =  —  ' 

Phantom  Circuit 

—  ^^n 

Wire  No.  3 

Phantom  Circuit 

Side  Circuit  f 

1 

_^_          «                Wire  No.  4 

•f] 

s 

, 
- 

3.   Side  Circuit 

3 

< —  instantaneous  direction  of  side-circuit  currents. 

< instantaneous  direction  of  phantom-circuit  currents. 

FIG.  25.     Phantom  Circuit  Derived  from  Two  Side  Circuits 

between  the  two  wires  of  each  pair,  causes  no  appreciable  induced  current  in  the  side- 
circuit  terminations.  Also,  the  side-circuit  currents  do  not  enter  the  phantom.  This  ar- 
rangement provides  three  circuits  from  four  wires  but  requires  a  well-maintained  balance 
in  the  line  sides  of  the  repeating  coils  and  between  the  four  wires,  and  adequate  trans- 
posing, both  in  the  open-wire  group  and  the  cable  quad,  to  prevent  excessive  noise  and 
crosstalk. 

17.  PLANT  DESIGN— EXCHANGE 

Exchange  facilities,  employed  in  connecting  subscriber  station  equipment  with  central 
offices,  for  interoffice  trunking  in  multioffice  exchange  areas,  and  for  miscellaneous  uses, 
may  consist  of  cable  or  open  wire  or  a  combination  of  both  types  of  facilities.  The  latest 


Gage  Cabl 

e 

26 

24 

22 

19 

16 

13 

Type  of  cable.  .  .  .  J 

ST 
AST 

BST 

M 
SM 
ASM 
GSM 

DSM 

NM 

SA 
ASA 
BSA 
CSA 

NA 
ANA 

TA 

TS 

BNB 
CNB 

TB 
ANB 
DNB 

TH 
NH 

TJ 

R,  ohms  per  mile  at 
68°  F  

440 
.069 

440 
.079 

274 
.072 

274 
.084 

274 
.065 

171 
.082 

171 
.073 

171 
.062 

171 
.068 

85 
.084 

85 
.066 

42 
.066 

21.4 
.066 

C,  Atf  per  mile  

Loading 

Load 
Spacing, 
feet 

Decibels  per  Mile  at  68  Deg  Fahr 

B-175 
B-I35 
B-88 

D-175 
D-135 
D-88 

H-250 
H-175 
E-135 
H-88 
E-44 

M-175 
M-135 
M-88 

R-133 

Non- 
loaded. 

3,000 
3,000 
3,000 

4,500 
4,500 
4,500 

6,000 
6,000 
6,000 
6,000 
6,000 

9,000 
9,000 
9,000 

11,600 

.94 
1.05 
1.30 

1.12 
1.25 
1.52 

.01 
.12 
.39 

.20 
.33 

1.62 

.63 
.69 
.87 

.74 
.82 
KOI 

.68 
.75 
.94 

.80 
.88 
1.09 

.44 
.48 

.25 
.26 

.22 
.24 

.14 
.14 

.60 

.51 
.56 
.70 

.49 

.34 

.28 
.30 
.38 

.30 

.25 
.27 
34 

.18 
.15 

.26 
.27 
.30 
.38 
.50 

.33 
.36 
.44 

.41 
1.11 

.17 
.15 
.16 
.21 
.27 

.17 
.20 
.24 

.21 
.75 

.11 
.10 

.11 

".\4 
.12 

.50 

.31 
.34 
.42 
.56 

1.40 
1.69 
2.06 

1.50 
1.80 
2.21 

.92 
1.14 
1.46 

.00 
.23 

.58 

.63 
.79 
1.04 

.60 
.65 

.60 

63 

1.63 
1.91 

1.75 
2.04 

1.09 
1.31 

.18 
.42 

1.14 
1.25 

.75 
.92 

.73 
.87 

.41 
.49 

76 

80 

2.67 

2.86 

2.14 

2.31 

2.04 

1.79 

1.69 

1.55 

1.63 

1.26 

PIG.  26.     Attenuation  Losses  of  Non-quadded  Exchange  Area  Cable  Facilities 


PLANT  DESIGN — EXCHANGE 


17-93 


developments  affecting  rural  service  include  (1)  trial  installations  of  telephone  carrier 
systems  on  rural  power  distribution  lines,  serving  areas  where  telephone  facilities  are  not 
available,  and  (2)  direct  radio  channels  between  the  central  office  and  farm  or  ranch  homes 
heretofore  economically  inaccessible  for  the  usual  pole  line  and  open-wire  construction. 
EXCHANGE  CABLES,  in  general,  employ  five  different  gages  of  soft-drawn  copper 
conductors,  namely,  26,  24,  22,  19,  and  16  gage.  A  relatively  small  amount  of  28-gage 
cable,  developed  to  conserve  copper  during  World  War  II,  has  been  manufactured,  but 
with  normal  copper  prices  its  use  in  place  of  26-gage,  where  applicable,  does  not  effect 
appreciable  cost  savings.  Exchange  trunk  cables  are  usually  of  19  or  smaller  gage. 


Type  of  Facility 

Values  Shown  Are 

R 
(Loop), 
ohms 
(dc) 

L. 

henrys 

<?, 
mhos 
(1000 

cycles) 
(X1Q-6) 

c, 

farads 
(X10~6) 

G 
C 

Per  Unit 
Length  of 

At 
Temp 
ofF° 

Dry 
or 

Wet 

Non-quadded  exchange  area  cables 

*"•    I            ST  % 

f  M     SM    ASM    GSM 
24  gage      {                              DSM 
I                              NM 
{SA    ASA     BSA     CSA 
NA    ANA 
TS 

19jra*e       I                     BNB     CNB 
iv  gage      {            TB    j^    I)NB 

16  gage                              TH      NH 
13  gage                                           TJ 

Mile 

68 
68 
68 
68 
68 
68 
68 
68 
68 
68 
68 
68 
68 

440 
440 
274 
274 
274 
171 
171 
171 
171 
85 
85 
42 
21.4 

0.001 
.001 
.001 
.001 
.001 
.001 
.001 
.001 
.001 
.001 
.001 
.001 
.001 

1.8 
2.1 
1.9 
2.2 
1.7 
2.1 
.9 
.6 
.7 
2.2 
.7 
.7 
.7 

0.069 
.079 
.072 
.084 
.065 
.082 
.073 
.062 
.068 
.084 
.066 
.066 
.066 

26 
26 
26 
26 
26 
26 
26 
26 
26 
26 
26 
26 
26 

Submarine  cables  —  non-quadded 
[24  gage 
Single  paper  insulation          s  22    u 
119    « 
f24gage 
1  "yy    K 
Double  paper  insulation        <**    u 

U6    * 

Mile 

55 
55 
55 
55 
55 
55 
55 

266 
166 
83 
266 
166 
83 
41 

.001 
.001 
.001 
.001 
.001 
.001 
.001 

.7 
.9 
2.0 
1.8 
2.1 
2.2 
1.7 

.066 
.075 
.078 
.071 
.080 
.083 
.066 

26 
26 
26 
26 
26 
26 
26 

17-gageUWire 
U  bridle  -wire 

_  .  \  distribution  wire  (buried) 

Eibfoot 
fEIofoot 

I     mile 

68 
68 

68 

Wet 

10.3 
10.3 

54 

,00033 
.00027 

.0014 

* 
7.6 
40.0 

.025 
f.023f 
1.026J 
f  J22t 

I.135t 

328  f 
2961 
328  f 
296* 

Drop  wires 

{g^ 
{&*? 

14  gage                                  EC  type 

Eolofoot 

68 
68 
68 
68 
68 

Wet 

51 
51 

28 
28 

5 

.00021 
.00023 
.00022 
.00022 
.00025 

* 
* 
* 

.042 
.036 
.040 
.040 
.041 

Miscellaneous  wires  and  cables 
Inside  wiring  cable  —  22  gauge 
f  OR  type 

TT>        « 

Service  cables—  22  gage      \^    u 
LR 

ITR  u 

AL  wire                                14  gage 
Bridle  wire                           20  gage 
Duct  wire                              22  gage  ) 
DTJ  station  wire                   22  gage  / 
GN  station  wire                   22  gage 

Kilofoot 

}; 

« 

u 

68 
68 

68 
68 

68 
68 

Wet 

37 
37  § 

5 
21 

33 
33 

.00020 
.  00027  § 

.00029 
.00028 

.  00030  B 
.  00030  f| 

* 
* 

* 
* 

* 
* 

,025 
.0205 

.033 
.036 

.033  |! 
.048  fi 

*  Leakage  conductance  at  1000  cycles  is  negligible  as  compared  with  capatitive  susceptance, 

t  Initial  values  after  one  day  soaking  in  water. 

j  Estimated  values  after  5  to  10  years  in  ground,  depending  upon  moisture  conditions  in  soil 

§  These  values  may  be  applied  to  both  one  and  two  pair  cables, 

j  These  values  are  satisfactory  for  pairs,  triples,  or  quads. 

FIG.  27.     Primary  Distributed  Constants  of  Cables 'and  Miscellaneous  Paired  Conductor  Facilities 


17-94 


TELEPHONY 


Cable 

Loading 

Propagation  Constant  at  68  Deg  Fahr 

Characteristic  Impedance  * 
at  68  Deg  Fahr 

Gage 

Code 

Per  Mile 

Per  Kilofoot 

26 

ST 
AST 

NL 
B-175 
B-135 
B-88 

,3072  +  ;  .3105 
.1084  +  ;'  .9354 
.1207+;   .8223 
.1492+;  .6713 

.05818  +;.  05881 
.  02053  +;M772 
.02286  +  /.1557 
.02826  +  ;.  1271 

718  -;706  =  1007  \~44755 
2204  -;251  -  2218rT"5* 
1929  -;28I  =  1949  r~O* 
1567  -;344  «  1604VT2T?5 

D-175 
D-135 
D-88 
H-135 

.1286+;  .7739 
.1434  +  ;  .6824 
.1747  +  ;   .5644 
.1615+;  .6030 

.  02436  +;M466 
.02716+;.  1292 
.03309  +;M069 
.03059  +  ;M142 

1848  -;299  -  1872\~Tr 
1618  -  ;332  =  1652  \TT765 
1325  -  ;403  =  1385^1679^ 
1440  -;383  «  1490VT4T9* 

H-88 
H-44 
M-135 
M-88 

.1940  +;'  .5049 
.2375+;  .4062 
.1880  +  ;  .5153 
.2196+;  .4424 

.03674  +;.  09563 
.04498  +;.  07693 
.03561  +  ;.  09759 
.04159  +;.  08379 

1192  -  ;453  =  1275\"20.86 
949  -;552  «  1  098  \  30.  2^ 
1257-;460  «  1338^0^ 
1057  -;525  «  IISOVIO5 

BST 

NL 
B-175 
B-135 
B-88 

.3287  +  ;  .3322 
.1160+;!.  0009 
.1292  +  ;   .8799 
.1596  +  ;  ,7183 

.06225+;.  06292 
.  02197  +;M896 
.02447+;.  1666 
.03023  +;.1360 

672-;660=     942V44T35 
2060  ~;235  «  2073  \    6^ 
1802  -;263  «  1821  \    8.  3* 
1464-;322  =  1499M2.45 

D-175 
D-135 
D-88 
H-135 

.1376+;  .8281 
.1534  +  ;  .7302 
.1869+;  .6039 
.1728+;  .6452 

.02606  +;M568 
.02905  +;.  1383 
.03540  +;M144 
.03273+;.  1222 

1727-;280  =  1750\"  9.2d 
1512~;310  «  1544\11,6° 
1238-;376  =  1294\16.9° 
1346»;358  =  1393\  14.9° 

H-88 
H-44 
M-135 
M-88 

.2076  +  ;  .5403 
.2541  +;  .4346 
.2012  +  ;  .5514 
.2350  +;  .4734 

.03932  +;.  1023 
.04813+;.  08231 
.03811  +;M044 
.04451  +;*.  08966 

1H4  -;423  =  1192\20.8° 
887  -  ;516  -  1026\30.2° 
1174  ~;430  =  1250\20.1° 
988  -  ;490  »  1103\26.4° 

24 

M 
SM 
ASM 
CSM 

NL 
B-175 
B-135 
B-88 

.2467+;  .2513 
.0722  +  ;   .9504 
.0794  +;'  .8344 
.  0998  +  3  .  6757 

.04672  +;.  04759 
.  01367  +j.  1800 
.01504  +;M580 
.01890  +;.  1280 

558-;542=     778  \  44.  2° 
2155  -;155  =  2161\    4.1° 
1880  -;171  =  1888\    5.2° 
1515  -  ;216  *  1530\    8.1° 

D-175 
D-135 
D-88 
H-135 

.0849+;  .7844 
.0941  +;  .6887 
.1165  +  ;  .5613 
.1063+;  .6035 

.01608+;.  1486 
.  01782  +  /.1304 
.  02206  +;M063 
.02013+;.!  143 

1800  -;186  =  1810\    5.9° 
1566-;209  =  1580\    7.6° 
1264-;257  =  1290\il.5° 
1386-;239  =  1407  \    9.8° 

H-88 
H-44 
M-135 
M-88 

.1309+;  .4945 
.1682  +;   .3763 
.1254  +  ;   .5066 
.1513+;   .4212 

.02479  +;.  09366 
.03185  +  ;.  07127 
.02375  +;.  09595 
.02866+;.  07977 

1123  -;292  «  1160\  14.6° 
844-;372=     922  \  23.  8° 
1187  -;294  =  1223\13.9° 
968  -;345  =  1028\  19.6° 

DSM 

NL 
B-175 
B-135 
B-88 

.2664+;  .2715 
.0780  +  ;  1.0266 
.0858  +;   .9013 
.1078+;  .7298 

.05045  +;.05142 
.01477+;.  1944 
.01625  +  ;.1707 
.02042+;.  1382 

517-;503«    721  \  44.  2° 
1996  -;143  =  2001  \    4.1° 
1741  -  ;158  -  1748\    5.2° 
1402  -  ;200  =  1416\    8.1° 

D-175 
D-135 
D-88 
H-135 

.0917  +;"  .8473 
.1016  +;   .7439 
.1258+;   .6063 
.1148+;  .6519 

.01737  +  ;.  1605 
.01924  +;M409 
,02383  +  ;.  1148 
.  02174  +;M235 

1667  -;172  »  I676\'5.9° 
1450  -;193  =  1463\    7.6° 
1170  -;238  »  1194\  11.5° 
1284  -;222  =  1303\    9.8° 

H-88 
H-44 
M-135 
M-88 

.1414  +;*  .5341 
.1817+;   .4065 
.1354+;   .5472 
.1634+;   .4550 

.02678  +;M012 
.03441  +;.  07699 
.  02564  +;M  036 
.03095  +  ;.  08617 

1039  -;271  «  1074\14.6° 
781-;345=     854  \  23.  8° 
1099-;272=  1132\  13.9° 
a97-;319=     952  \  19.  6° 

NM 

NL 

.2342  +;'  .2388 

.04436  +  ;.  04522 

588  -  ;572  -     820  \44.2° 

*  Mid-section  iterative  impedance  in  cases  of  loaded  facilities. 
FIG.  28.     Secondary  Constants  of  Exchange  Area  Cable  Facilities  at  1000  Cycles  per  Second — 26  and 


PLANT  DESIGN — EXCHANGE 


17-95 


Cable 

Loading 

Propagation  Constant  at  68  De&  Fahr 

Characteristic  Impedance  * 
at  68  Deg  Fahr 

Gage 

Code 

Per  Mile 

Per  Kilofoot 

22 

SA 
ASA 
BSA 
CSA 

NL 
B-175 
B-135 
B-88 

.2065+y  .2134 
.  0503  +yi.  0155 
.0549  +  y  .8900 
.0689+y  .7177 

.03911  +y.  04042 
.  00953  +y.  1923 
.01040  +  j.l  686 
.  01305  +y.  1359 

416-y399~    576  \  43.  8° 
2025  -j  92  =»  2027  \    2.6° 
1762-jl02»  1765V  3.  3° 
1414  -yi30  =  1420  \    5.3° 

D-I75 
D-I35 
D-88 
H-I35 

.0583+y  .8365 
.0647+j  .7325 
.0808+j  .5922 
,0729+y  .6402 

.  01  104  +y.  1584 
.01  225+  j.l  387 
.  01530  +J.1122 
.01381  +y.!213 

1694-yil3«  1698V  3.8° 
1465  -  yi25  =  1470\    4,9° 
1170  -yi56-  IISOV'O5 
1298-yi44«  1306\    6.3° 

H-88 
H-44 
M-135 

M-88 

.0907+y  .5185 
.1199+y  .3796 
.0863+y  .5333 
.1060+j   .4341 

.  01718  +J.09820 
.02271  +y.  07189 
.  01634  +J.1010 
.  02008  +y.  08222 

1036-  J177-  1051  rO3 
748-y233  =    783  \  17.  3° 
1109  -yi78-  1123\    9.1° 
879-j214=    905\13.7° 

NA 
ANA 

NL 

.1946  +  j  .2012 

.03686  +y.  03811 

442-j424=    612  HO5 

TA 

NL 

.1792+j  .1853 

.  03394  +j.  03509 

479-j460-    664  \  43.  8° 

TS 

NL 

.1882  +j  .1945 

.  03564  +j.  03684 

457  _  j438  =    633  \43.8° 

19 

BNB 

CNB 

NL 
B-135 
B-88 

.1446+y  .1551 
,0304+j  .900 
.0386+j  .725 

.02739+;.  02938 
.00576+  j.l  705 
.00731  +  J.1373 

295-j273=    402  \  42.  8° 
1741  -j  52=  1742\    1.7° 
1393  -j  69-  1395\   2.8° 

D-175 
D-135 
D-88 
H-135 

.0321  +y  .8457 
.0349+j  .740 
.0439+j  .5957 
.0388  +y  .6455 

.  00608  +j.  1602 
.00661  +  j.  1402 
.00831  +j.  1128 
.  00735  +y,  1223 

1676  -j  58  -  1677\   2.0° 
1448  -j  63*  I449\   2.5° 
1155  -j  81  =  I158\   4.0° 
1281  -j  74-  I283\    3.3° 

H-88 
H-44 

M-88 

.0487+j  .5194 
.0645+j  .3701 
.0568+j  .4302 

.  00922  +y.  09837 
.  01222  +j.  07009 
.  01  076  +j.  08148 

1013  ~y  92=  1017\    5.2° 
713  -j  122=    723\'  9-7° 

854  _  .fin  .  861  rr^ 

TB 
ANB 
DNB 

NL 
B-175 
B-135 
B-88 

.1282+y  .1375 
.0254+y  .908 
.0270+y  .795 
.0342+j  .641 

.  02428  +j.  02604 
.00481  +y.  1720 
.00511  +  /.1506 
.  00648  +J.1214 

333-j308=    453  V4O5 
2237  -y  54=  2238  FT?5 
1951  _  j  61  =  1952\    1.8° 
1563  -j  76=  1565  \~2TF 

D-175 
D-135 
D-88 
H-175 

.0282+y  .7461 
.0310  +j  .653 
.0390  +  y  .5269 
.0315+y  .6507 

.  00534  +j.  141  3 
.00587+  j.l  237 
.  00739  +/.  09979 
.00597  +  J.1232 

1862  —  j  65  =  1863  \    2.0° 
1618  -y  71  *  1620\   2.5° 
1292  -j'  91  =  1295^^0* 
1643  -j  75  =  1645\   2.6° 

H-135 
H-88 
H-44 
M-88 

.0345+y  .5694 
.0432+j  .4590 
.0571  +y  .3282 
.0505+y  .3796 

.00653  +  J.1078 
.  0081  8  +j.  08693 
.01081  +  J.06216 
.  00956  +j.  071  89 

1423  -j  82=  1425  \   3.3d 
1132  -jl03  =  1137\    5.2° 
799-J138  =  .814  \   9.8° 
948-j'123=    956\   7.4° 

16 

TH 

NN 

NL 
B-175 
B-135 

.0868  +y  .1004 
.0156  +j  .908 
.0158+j  .795 

.  01644  +J.01902 
.  00295  +j.  1720 
.00299+  j.l  506 

243-j"208=    320\40.6° 
2238  -  y  30  -  2238  \   0.8° 
1951  -y  31  -  1951  \   0.9° 

B-88 
D-175 
H-175 
H-135 

.0203+j  .641 
.0168+j  .765 
.0178  +j  .6503 
.0188  +j  .5687 

.  00384  +y.  121  4 
.  00318  +j.  1449 
.00337  +  j.l  232 
.00356+  j.l  077 

1564  -j  44=  1565  \    1.6° 
1824  -j  64=  1825\    2.0° 
1648  -y  41  =  I649rT45 
1419  -y  42=  1420  VI.  7° 

H-88 
H-44 
M-88 

.0238+y  .4577 
.0307+j  .3249 
.0271  +j  .3773 

.00451  +j.  08669 
.00581  +/.Q6I53 
.00513  +J.07146 

1129  -j  55  =  1130  \   2.8° 
791  -y  72=*    794  \   5.2° 
934  -j  75=    937  \   4.6° 

*  Mid-section  iterative  impedance  in  case  of  loaded  facilities. 


FIG.  29. 


Secondary  Constants  of  Exchange  Area  Cable  Facilities  at  1000  Cycles  per  Second — 22, 
and  16  Gage 


17-96 


TELEPHONY 


Exchange  cables  are  usually  of  the  non-quadded,  paper-  or  pulp-insulated  types,  having 
attenuation?  losses  at  1000  cycles,  as  shown  in  Fig.  26.  These  cables  are  not  generally 
loaded  for  subscriber  loop  use,  but  are  frequently  loaded  when  employed  for  interoffice 
trunks,  particularly  in  the  larger  exchange  areas.  The  primary  distributed  constants  of 
exchange  area  cable  facilities  and  of  miscellaneous  paired  conductors  for  exchange  use  are 
shown  in  Fig.  27.  The  resistances,  inductances  and  capacitances  are  in  d-c  values,  but 
for  practical  purposes,  may  be  considered  equivalent  to  the  1000-cycle  values.  The 
leakage  conductances  are  specifically  1000-cycle  values.  The  secondary  constants  of 

of  exchange   area   cable    facilities    at    1000 
cycles,  are  shown  in  Figs.  28  and  29. 

OPEN- WIRE  FACILITIES  for  exchange 
use  are  principally  of  iron  or  high-tensile- 
strength  steel,  the  steel  being  used  almost 
exclusively  in  recent  years,  owing  principally 
to  economy  and  better  service  performance. 
New  types  of  steel  have  been  developed  for 
telephone  wire  as  the  result  of  extensive 
studies  of  the  materials  used,  and  by  means 
of  heat  and  other  treatments  applied  during 
manufacture.  Some  buried  wire  (such  as 
W.E.  Co.  U  or  UA  types,  loaded  or  non- 
loaded)  is  used  in  rural  areas  for  short  dis- 
tances, the  characteristics  of  which  are  given 
in  Fig.  27. 

The  a-c  resistance  of  BB  grade  iron  wire 
and  of  Crapo  HTL-85  and  HTL-135  high- 
tensile  telephone  line  wire,  over  a  frequency 
range  of  500  to  2000  cycles,  using  current  of 


CC  I! 

s 


S 

S* 

oc 


500  750        1000        1250       15  OO       1750     2000 

FREQUENCY  IN  CYCLES  PER  SECOND 

Fia.  30.     Comparison  of  A-c  Resistance  of  BB 


tto^Ll^tl^l^l^WoS?    J"^?  magnitude,  i.  shown  in  Fig.  30. 
1945  by  Indiana  Steel  and  Wire  Co.)  The  smaller  variation  in  a-c  resistance  of  the 

Crapo  wire  (manufactured  by  Indiana  Steel 

and  Wire  Co.),  as  compared  to  the  BB  wire,  tends  to  reduce  modulation  effects  on  voice 
currents  traveling  over  the  wire,  and  thus  to  improve  the  quality  of  the  transmitted 
speech.  Figure  31  shows  comparative  breaking  strengths  between  the  above  three  wires, 
as  presented  by  Indiana  Steel  and  Wire  Co.  Figure  32  shows  the  characteristics  at  68 
deg  fahr  of  109  (No.  12  BWG)  high-strength  steel  and  134  (No.  10  BWG)  steel  wire  with 
0.8-oz  zinc  coating,  12-in.  spacing,  and  DP  insulators. 

Copper  steel  wire  (104  mil  diameter),  the  characteristics  of  which  are  shown  in  Table 
5,  has  roughly  one-half  the  attenuation  per  mile  at  1000  cycles  of  109  high-strength  steel 
wire  (Fig.  32).  Thus,  this  wire  may  be  used  advantageously  in  place  of  high-strength 
steel  wire  where  the  lower  loss  is  required  to  meet  exchange  transmission  standards.  Cop- 
per steel  wire  (0.081  mil  diameter  and  40  per  cent  conductivity)  having  per  pair  mile  an 
attenuation  loss  of  0.22  db  (wet)  and  resist- 
ance of  42.8  ohms  at  68  deg  fahr  may  also 
be  used  where  applicable. 

Buried  wire  (paired,  insulated,  such  as 
W.E.  Co.  U  or  UA  type)  is  sometimes  used, 
loaded  or  non-loaded,  in  place  of  or  as  an  ex- 
tension of  open-wire  rural  plant,  depending 
on  economies. 

CARRIER  CHANNELS  superimposed  on 
rural  power  lines  are  now  under  trial  opera- 
tion to  determine  the  practicability  and  re- 
quirements for  thus  serving  rural  subscribers 
in  locations  not  at  present  served  by  tele- 


HTL 
85 


BB 


>  200       400        600        800       1000        1200 

BREAKING  STRENGTH  IN  POUNDS 

31.  Comparative  Breaking  Strength  of  BB 
Grade  Iron  Wire  and  Two  Types  of  High  Tensile 
Telephone  Line  Wire — No.  12  BWG  (Copyright 
by  Indiana  Steel  and  Wire  Co.) 


phone  lines.  The  equipment,  developed  by  the  Bell  System  and  designated  as  the  M-l 
carrier  telephone  system,  is  being  made  in  quantities  by  the  Western  Electric  Co.,  the 
privilege  of  producing  it  being  extended  to  other  manufacturers. 

The  first  two  systems,  installed  in  1945  for  trial  at  Jonesboro,  Axk.,  and  Selma,  Ala., 
provide  rural  service  to  four  subscribers  over  an  11-mile  section  of  7200-volt,  multi- 
grounded  neutral,  single-phase  power  line  out  of  Jonesboro,  and  to  four  subscribers  over 
a  14-mile  section  of  6900-volt,  multigrounded  neutral,  single-phase  power  line  out  of 
Selma, 

The  principal  elements  of  the  rural  power  line  carrier  system,  as  now  developed,  are 
shown  in  Fig.  33.  Single  channel  operation  is  employed,  using  for  the  trials  frequencies 


PLANT  DESIGN — EXCHANGE 


17-97 


0.8-oz.  Zinc  Coating,  12-in.  Spacing,  68  Deg  Fahr 
DP  Insulators 
109  Side  Circuits 


Fre- 
quency, 
ko 

Attenuation, 
db/mi 

Phase  Shift, 
radians/mi 

Characteristic  Impedance 

Dry 

Wet 

Dry 

Wet 

Dry 

Wet 

0.3 
1.0 
2.0 
3.0 
4.0 

.181 
.289 
.407 
.528 
.623 

.207 
.313 
.436 
.560 
.660 

.028 
.067 
.121 
.167 
.212 

.026 
.068 
.095 
.169 
.215 

1754  -  yi314  =  21  92  \  36.  83° 
1279  -y  629  =  1425  \26.1  8° 
1142  -y  443  =  1  225  \  21.20° 
1054  -  j  383  =  1121  \  19.97° 
1004  -  j  339  =  1060  \  18.65° 

1818-^1122  =  2I36\31.68° 
1274  -  j  578  =  1  399  \  24.  40° 
1130  -y  413  *  1203  \20.0L6 
1042  -j  360  »  1102\  19.07° 
991  -y  320  =  104I\  17.90° 

109  Phantoms  (of  Non-pole  Pairs) 


0.3 
1.0 
2.0 
3.0 
4.0 

.166 
.259 
.366 
.466 
.548 

.195 
.286 
.396 
.502 
.588 

.026 
.064 
.115 
.160 
.204 

.025 
.065 
.116 
.162 
.207 

969  -  y  713  »  1  203  \  36.  35° 
721  -y  334=     795  \  24.  85° 
642  -y  237=     684  \  20.  27° 
598  -  y  200  =     631  \  18.50° 
572  -j   176=    598\  17.10° 

1006  -y  588  =  1  165  \  30.  30° 
717  -j  300  =     777\22.70° 
635  -j  215  -     670  \  18.70° 
589  -y  185  =     617\  17.43° 
563  -j   164  =     586  \  1  6.  23° 

184  Side  Circuits 

0.3 
1.0 
2.0 
3.0 
4.0 

.136 
.250 
.389 
.502 
.599 

.159 
.273 
.416 
.533 
.632 

.026 
.069 
.120 
.167 
.209 

.025 
.070 
.121 
.168 
.212 

1563  -  y  945  =  1826  \  31.  17° 
1252  -  j  520  =  1356\22.55° 
1084  -y  404  =  1157\20.43° 
1004  -  j  348  =  1063\19.12° 
945  -j  311  =    995  \  18.22° 

1599  -y  788  =  1  783  \  26.  23° 
1245  -y  473  =  1  332  \  20.  80° 
1073  -y  377  =  1137X19.36° 
992  -  j  328  =  1045\  18.30° 
933  -j  294  =    978  \  17.  48° 

134  Phantom  Circuits  (of  Non-pole  Pairs) 

0.3 
1.0 
2.0 
3.0 
4.0 

.123 
.220 
.341 
.436 
.516 

.149 
.246 
.371 
.470 
.554 

.024 
.064 
.112 
.156 
.197 

.023 
.064 
.114 
.158 
.200 

883  -y  524  =  1  027  \  30.  68° 
711  -j  281  =    765  \  21.  57° 
625  -y  219  =     662\19.32° 
579  -y  185  =    608  \  17.  72° 
548  -y  165=    572  \  16.  75° 

902  -y  415  «    993  \  24.  70° 
706  -j  249  =    749  \  19.  43° 
617  -y  199  =    648  \  17.88° 
570  -j  172*    595  \  16.79° 
539  -y  153  =    560  \  15.85° 

FIG.  32.     Characteristics  of  109  (No.  12  BWG)  High-strength  Steel  and  134  (No.  10  BWG)  Steel  Wire 


COUPLING                                          POWER  -DISTRIBUTION 
CAPACITOR                                                TRANSFORMER 

FUIE         /  POWER  PHASE  WIRE  (70OO  VOLTS)         {              _      f/SE 

ISOLATING^ 
CHOKE   "H^ 

JJ^AJi^ 

-J     H            POWER  NEUTRAL  WIRE 
^V                       (GROUNDED) 

r°n  ' 

JS           TO 
=r)       OTHER 
yV  STATIONS 

TRUNKS  
OR  TOLL.. 
CIRCUITS 

TELEPHONE 
CARRIER  DROP 

*/^WtRING  INSTALLED  BY 
/*•'      POWER  LINEMAN 

1<-LINE  COUPLING  UNIT 
J     (CONTAINING  DRAINAGE 
j      INDUCTANCE  FILTER  AND 

p 

J; 

<-• 

- 

^  * 

'     COUPLING 
CAPACITOR 

LINE 
—  COUPLING 
UNIT 

*  —  POLE 
GROUND  WIRE 

TELEPHONE 
—  CARRIER 
DROP 

USUAL 
—  STATION 
PROTECTION 

INSIDE 
WIRING 

1  

120 
VOLT 
—      60 
—  CYCLE 
POWER 
SUPPLY 

PROTECTOR) 
POWER  METER—  *T 

HOUSE  LIGHTS     
AND  APPLIANCES  

60  CYCLE 
POWER  FOR  CARRIER  > 
STATION  SET 

COMMON 
CARRIER 
TERMINAL 
SERVING 
UP  TO  8 
PARTIES', 
POLE   OR 
OFFICE 
MOUNTED 

VOICE- 
D-FREQUENCY 
CIRCUIT 

TELEPHONE  ^B^ 

SET         tea 

j 

USUAL  VOICE 
FREQUENCY 
SWITCHING 
EQUIPMENT 
AT 
CENTRAL 
OFFICE 

POLE-> 
GROUND 
WIRE 

SUBSCRIBER         { 
CARRIER  TERMINAL  i 
,_             1 

GRC 

JUNID                               RURAL  RESIDENCE                 GROUND 

FIG.  33.    Principal  Elements  of  Rural  Power  Line  Carrier  Telephone  System  (Courtesy  Bell  System) 


17-98 


TELEPHONY 


of  165  kc  central  office  calling  subscriber  and  subscriber  receiving,  195  kc  subscriber  call- 
ing central  office  and  subscriber  talking  out  through  central  office,  and  185  kc  one  sub- 
scriber calling  another  subscriber  on  the  same  line  but  through  the  central-office  carrier 
terminal  equipment.  The  M-l  system  is  designed,  however,  for  double  sideband  carrier 
transmitted  amplitude  modulation,  with  as  many  as  six  channels,  each  serving  eight  sub- 
scribers and  each  using  three  frequencies.  These  frequencies  are  different  within  each 
channel  and  for  each  channel,  being  selected  for  transmission  from  the  common  terminal 
to  the  stations  within  the  range  155  to  230  kc  and  for  transmission  from  the  stations  to 
the  common  terminal  within  the  range  290  to  450  kc. 

Figures  34  and  35  show  block  diagrams  of  the  common  carrier  terminal  at  the  central- 
office  end  of  the  system  and  of  the  subscriber  carrier  terminal  (station  set) ,  respectively, 
of  the  types  used  in  the  trial  tests. 


RECEIVING 
AMPLIFIER 

2 

— 

195-KC 
FILTER 

FIG.  34.    Block  Diagram  of  Common  Carrier  Terminal  at  the  Central-office  End  of  a  Rural  Power  Line 
Carrier  Telephone  System  (Courtesy  Bell  System) 

The  carrier  terminals  are  connected  to  the  power  line  through  a  0.002-juf  capacitor 
(8700-volt  rating  for  the  trials),  and  a  line  coupling  unit.  This  unit  has  a  drainage  in- 
ductance coil,  filter,  and  protector.  All  branches  of  the  power  network  not  used  for  car- 
rier transmission  have  inserted  in  the  primary  phase  conductor  an  isolating  choke  for  the 
main  power  circuit  and  a  tap  choke  for  branches  from  the  main  circuit.  Transmission 
chokes  are  used  in  branches  from  the  main  power  circuit,  where  the  branch  is  being  used 
for  carrier,  in  order  to  reduce  the  bridging  loss  of  the  branch  to  through  transmission  over 
the  main  line.  The  common  carrier  terminal  is  designed  to  terminate  the  power  line  in 
about  500  ohms  impedance  for  the  carrier  frequencies  used,  as  are  the  coupling  capacitors 
and  line  coupling  units  at  subscriber  stations  or  at  the. end  of  a  power  line  tap  where  no 
subscriber  station  exists. 

Each  carrier  terminal  requires  continuous  110-120  volt,  60-cycle  a-c  power  supply,  and 
the  unmodulated  carrier  power  delivered  to  the  line  coupling  unit  from  each  terminal  is 
about  1  watt.  The  subscriber  terminal  requires  about  8  watts  of  standby  and  30  watts 
of  operating  power. 

The  common  carrier  terminal  connects  to  the  manual  switchboard  or  dial  unit  over  a 
regular  two-wire  voice  frequency  circuit.  Divided  code  ringing  is  provided  using  inter- 
rupted carrier  current  in  proper  time  sequence,  or  bridged  code  ringing  can  be  used  if 
required. 


PLANT  DESIGN — EXCHANGE 


17-99 


Where  the  power  circuit  is  not  of  the  single-phase,  multigrounded  neutral  type  or  it  is 
desired  to  operate  more  than  one  system  on  the  same  power  line  but  in  different  sections, 
special  consideration  of  the  factors  involved  will  be  required. 

Preliminary  data  indicate  that  the  average  transmission  loss  between  the  common  ter- 
minal and  the  most  distant  station  at  the  operating  carrier  frequencies  of  the  M-l  system, 
including  subscriber  coupling  unit  bridging  and  transmission  choke  losses,  will  be  about 
2.0  to  2.5  db  per  mile.  The  overall  loss  between  the  common  terminal  and  any  station 
should  not  exceed  35  to  40  db  (15  to  18  miles  of  line).  Under  low  atmospheric  static  con- 
ditions, the  carrier  system  can  be  adjusted  to  provide  about  the  same  effective  transmission 
(T  •+-  R)/2  from  any  station  to  the  central  office  as  would  be  provided  from  a  regular 
voice-frequency  station  having  the  latest-type  antisidetone,  local  battery  talking  set 
(W.B.  Co.  FIA-AST-LBT-2  cells). 

The  M-l  carrier  system  may  also  be  adapted  to  telephone  wire  lines,  where  additional 
telephone  circuits  may  thus  be  provided  more  economically  than  by  other  methods.  This 
usage  requires  further  study. 


RECEIVER 


FIG.  35.    Block  Diagram  of  Subscriber  Station  Set  of  a  Rural  Power  Line  Carrier  Telephone  System 

(Courtesy  Bell  System) 

In  operation,  the  system  responds  as  though  the  connecting  circuit  between  the  central 
office  and  subscriber  were  the  usual  wire  line.  On  revertive  calls  (one  subscriber  calling 
another  on  the  same  circuit  or  channel),  the  calling  subscriber  places  the  call  to  the  called 
subscriber  in  the  usual  manner  (through  the  operator  or  equipment)  and  then  puts  his 
handset  on  the  cradle.  When  the  ringing  signal,  which  the  calling  subscriber  hearsT  is 
stopped,  indicating  that  the  called  subscriber  has  answered,  the  calling  subscriber  removes 
his  handset  from  the  cradle  and  talks.  The  conversation  is  carried  on  through  the  com- 
mon terminal,  and  operator  supervision  is  provided  if  the  office  is  of  the  manual  type. 

Radio  channels  for  rural  telephone  service  are  under  trial  test,  as  discussed  in  article  8 
of  this  section.  It  is  expected  that  suitable  equipment  will  be  developed,  based  on  these 
tests,  which  will  permit  utilizing  radio  channels  for  this  service  on  a  commercial  basis. 

SUBSCRIBER  LOOP  DESIGN.  The  design  procedure  and  considerations  in  engineer- 
ing subscriber  loop  plant  are  briefly  summarized  (for  a  single  cable  route)  as  follows: 

(a)  Determine  the  most  economical  gage  requirements  for  the  ultimate  area  to  be  served 
by  the  various  complements  of  cable,  considering  both  transmission  and  signaling  design 
limits,  using  the  effective  subscriber  loop  loss  curves,  if  available. 

(6)  Examine  the  possibilities  of  obtaining  further  cable  economies  by  reducing  bridged 
tap  or  distribution  cable  losses  which  were  assumed  in  determining  the  design  limits. 

(c)  Examine  the  possibilities  of  utilizing  any  existing  plant  with  the  new  plant  being 
designed,  without  exceeding  the  design  limits. 

(d)  If  the  loop  loss  or  signaling  limits  are  exceeded  in  existing  plant,  determine  the 
most  economical  plan  of  meeting  the  limits,  such  as  using  special  sets,  reducing  bridged 
cable  losses,  loading  the  longer  cable  pairs,  or  choosing  larger-gage  cable  or  long  line 
circuits. 


17-100  TELEPHONY 

(e)  If  some  margin  of  loop  loss  and  loop  resistance  results  from  the  use  of  existing  plant 
in  connection  with  the  plant  being  designed,  study  the  advisability  of  smaller-gage  pairs 
than  indicated  under  (a)  above,  taking  into  consideration  the  permanence  of  the  existing 
facilities  and  the  relation  of  the  proposed  plant  to  the  ultimate  plan. 

(/)  Combinations  of  gages  are  frequently  practicable,  particularly  since  the  trend  is 
toward  the  finer-gage  cable  adjacent  to  the  central  office. 

(g)  Composite  (more  than  one  gage)  cables  are  installed  extensively,  the  larger  gages 
serving  the  outlying  areas. 

(A)  Loading  on  long  loops  is  sometimes  employed  to  meet  loop  loss  limits.  H-44  or 
H-88  loading  on  19-,  22-,  or  24-gage  cable  results  in  substantial  loss  reductions  over  non- 
loaded  loops,  which  might  permit  smaller-gage  conductors  for  long  loops.  Bridged  taps 
on  loaded  pairs  must  be  limited  to  avoid  serious  impairments. 

(i)  Signaling  limits  may  economically  be  extended,  in  some  cases  by  modifications  or 
readjustments  of  central-office  equipment,  larger-gage  conductors  or  long  line  units  for 
the  longer  loops,  or  occasionally  the  use  of  two  pairs  in  parallel  may  be  justified  to  reduce 
loop  resistance. 

0')  Loop  and  trunk  plant  design  is  based  on  the  most  efficient  types  of  subscriber  sets 
available  or,  in  some  cases,  anticipated  within  a  relatively  short  period  of  time. 

(fc)  Special  exchange  lines,  such  as  private  or  PBX  tie  and  foreign  exchange  lines,  con- 
ference and  bridging  arrangements,  and  one-way  speech  networks,  require  special  design 
work  to  meet  their  particular  needs. 

The  zoning  of  subscriber  sets  is  a  procedure  that  provides  for  the  introduction  of  station 
apparatus  in  such  a  way  as  to  obtain  the  desired  grade  of  transmission  at  the  lowest  prac- 
ticable overall  cost.  Owing  to  the  different  efficiencies  of  the  various  types  of  station  ap- 
paratus in  service  and  under  continuous  development,  it  is  necessary  to  insure  that  this 
apparatus  will  be  installed  as  required,  so  that  (1)  the  cost  of  the  outside  plant  will  be  the 
minimum,  (2)  transmission  will  be  satisfactory  for  the  outside  plant  design,  and  (3)  station 
apparatus  costs  will  be  minimized  by  obtaining  a  reasonable  service  life  for  the  older  ap- 
paratus, avoiding  as  far  as  possible  premature  replacements  of  existing  apparatus  and 
providing  an  orderly  program  for  the  introduction  of  new  station  apparatus.  Thus,  ex- 
change areas,  as  required,  are  divided  into  zones  within  which  only  certain  types  of  station 
apparatus  may  be  used. 

TRUNK  PLANT  DESIGN  is  based  on  the  transmission  loss  limit  assigned  to  each  group 
of  trunks,  as  determined  from  the  loop  and  trunk  study.  The  design  limit  is  obtained  by 
deducting  from  the  overall  permissible  trunk  loss  (1)  terminal  junction  losses,  due  to  char- 
acteristics of  the  proposed  trunks,  which  differ  from  the  reference  trunk;  (2)  intermediate 
junction  losses,  due  to  a  trunk  being  composed  of  different  types  of  facilities;  (3)  losses 
due  to  loading  irregularities;  and  (4)  equipment  and  office  losses  at  intermediate  points 
between  the  trunk  terminals.  Data  have  been  prepared  for  the  Bell  System  showing 
both  terminal  and  intermediate  junction  losses  under  the  usual  conditions  encountered 
in  practice.  In  some  cases,  under  certain  conditions,  these  losses  are  actually  negative 
(transmission  gains). 

The  design  limits  for  the  trunks,  having  been  determined  for  the  various  conditions 
under  which  the  given  group  of  trunks  will  operate,  represent  the  permissible  effective 
trunk  losses.  The  type  of  trunk  is  then  selected  which  meets  the  design  limits  and  has 
least  outside  plant  costs,  taking  into  consideration  existing  trunk  plant,  future  trunk  fa- 
cility needs  over  the  route  involved,  and  any  other  factors  that  may  have  a  bearing  on  the 
selection  of  the  type  of  trunk.  Figure  36  shows  effective  losses  for  non-quadded  exchange 
area  cable  trunks. 

The  signaling  limits  for  trunks  are  based  on  signaling,  pulsing,  and  supervision  require- 
ments, which  vary  materially  for  different  types  of  offices  and  associated  terminal  equip- 
ment. These  limits,  as  well  as  leakage  requirements,  are  usually  indicated  on  the  stand- 
ard drawings  for  each  type  of  central-office  circuit,  which  may  be  different  for  the  two  ends 
of  the  given  trunk.  The  lower  value,  if  there  is  a  difference,  will  be  controlling  in  deter- 
mining the  signaling  limit. 

Overall  trunk  resistance  for  signaling  purposes  is  usually  computed  from  unit  values 
for  temperatures  of  68  deg  fahr.  For  underground  trunk  plant  the  resistance  change  with 
temperature  will  be  about  3  per  cent  maximum,  which  may  usually  be  disregarded.  For 
aerial  cable,  the  change  may  be  as  much  as  10  to  12  per  cent,  and  this  variation  of  resist- 
ance with  temperature  change  should  be  considered  in  the  signaling  design  limits. 

With  the  larger-gage  trunk  facilities  generally  used  in  the  past,  the  transmission  design 
limit  usually  controlled  the  selection  of  the  type  of  trunk  facilities,  but  with  the  higher 
permissible  interlocal  trunk  losses  in  recent  years  (resulting  from  allocation  to  trunks  of 


PROGRAM  SERVICE 


17-101 


part  of  the  new  instrument  gains)  and  the  trend  toward  higher  supervision  limits  with 
current  types  of  dial  equipment,  the  field  of  use  of  the  finer-gage  cables  has  been  extended, 
so  that  signaling  limits  may  be  controlling  in  some  cases. 


Cable                   { 

26 
AST 

24 

ASM 

22 
BSA 

19 
CNB 

19 
DNB 

16 
TH 

13 

TJ     • 

Load- 
ing 

Spacing 

ft 

Decibels  per  Mile  at  68  Beg  Fahr 

NL 

3.20 

2.57 

2.15 

1.51 

1,34 

0.91 

0.60 

M-88 
M-135 
M-175 

9000 
9000 
9000 

1.35 
1.22 

0.95 
0.79 

0,51 
0.42 

0.46 
0.37 
0.34 

.26 
.21 
.17 

.16 

.12 

H-44 
H-88 
H-135 
H-175 

6000 
6000 
6000 
6000 

1.48 
1.15 

1.05 
0.80 
0.64 

0.57 
0.43 
0.35 
0.32 

0.51 
0.39 
0.31 
0.28 

.27 
.21 
.17 
.16 

.11 

D-175 

4500 

0.52 

0.29 

0.26 

.15 

B-88 
B-135 
B-175 

3000 
3000 
3000 

0.87 

0.61 
0.49 

0.35 
0.27 
0.26 

0.31 
0.25 
0.23 

.18 
.14 
.14 

Decibels  per  Mile  at  110  Deg  Fahr 

NL 

3.33 

2.68 

2.24 

1.57 

1.39 

0.94 

0.63 

M-88 
M-135 
M-175 

9000 
9000 
9000 

1.47 
1.33 

1.03 
0.86 

0.56 
0.46 

0.50 
0.40 
0.37 

.28 
.23 
.18 

.17 

.13 

H-44 
H-88 
H-135 
H-175 

6000 
6000 
6000 
6000 

1.61 
1.25 

1.15 
0.87 
0.70 

0.62 
0.47 
0.38 
0.35 

0.55 
0.42 
0.34 
0.31 

.30 
.22 
.19 
.18 

.12 

D-175 

4500 

0.56 

0.31 

0.28 

.17 

B-88 
B-135 
B-175 

3000 
3000 
3000 

0.94 

0.66 
0.53 

0.38 
0.30 
0.28 

0.33 
0.27 
0.25 

.20 
.15 
.15 

Nate:  The  resultant  trunk  loss  should  not  be  used  closer  tiian  to  nearest  0.1  db. 
FIG.  36.     Effective  Trunk  Losses — Exchange  Area  Cable  Trunks 


PROGRAM  SERVICE 
18.  PROGRAM  SERVICE 

In  general,  program  service,  as  furnished  by  the  telephone  companies,  consists  mainly 
of  providing  suitable  wire  or  carrier  facilities  to  the  broadcasting  companies  for  the  trans- 
mission of  program  material,  which  usually  originates  in  broadcasting  studios  or  other 
locations  and  which  is  broadcast  from  radio  transmitters  to  the  public. 

Programs,  being  of  a  varied  nature,  from  the  finest  orchestral  music  to  ordinary  speech, 
require  different  grades  of  telephone  facilities  to  meet  the  broadcaster's  requirements.  For 
this  reason  the  telephone  companies  have  developed  and  have  made  available  for  broad- 
casting purposes  several  rather  broad  classifications  of  facilities,  as  shown  in  Table  1. 

SERVICE  REQUIREMENTS,  being  more  exacting  for  high-quality  than  the  lower- 
quality  program  circuits,  involve  the  control  of  (1)  transmission  levels  and  losses,  (2) 
frequency,  delay,  and  phase  distortion,  and  (3)  noise  and  crosstalk.  See  Section  12  for 
critical  cutoff  points  in  the  frequency  ranges  of  various  musical  instruments  and  speech. 

Transmission  levels  and  losses  in  open  wire  and  cable  are  maintained  as  specified,  by 
special  program  amplifiers,  the  latest  types  having  a  flat  gain  characteristic  with  a  max- 
imum gain  of  about  30  to  40  db  and  outputs  of  about  10  to  20  db  above  reference  volume, 


17-102 


TELEPHONY 


PROGBAM  SERVICE 


17-103 


depending  on  the  type  of  amplifier.  The  levels  must  be  high  enough  to  provide  a  satis- 
factory signal-to-noise  ratio  and  must  also  be  coordinated  with  the  levels  of  adjacent  cir- 
cuits or  systems,  to  avoid  causing  interference  in  them.  A  variable  attenuator,  having  a 
range  of  0  to  32  dfy  controls  the  amplifier  gain.  Figure  1  shows  a  typical  layout  and 
level  diagram  for  a  wide-band  (8000  cycles)  open-wire  program  system. 

Attenuation  variations  in  the  line  over  the  frequency  band  (frequency  distortion)  are 
compensated  for  by  low-  and  high-frequency  equalizers  of  the  constant-resistance  type, 
which  correct  for  variations  in  the  preceding  line  section,  to  meet  requirements. 

A  repeating  coil  (impedance  ratio,  line  to  drop,  of  1  :  1.15)  is  provided  at  each  repeater 
point  between  the  incoming  line  and  line  equalizing  apparatus,  which  latter  is  on  the  line 
side  of  the  line  amplifier.  This  coil  gives  the  proper  termination  to  the  line  and  insulates 
the  line  from  the  terminal  equipment  against  noise  and  for  protection.  Suitable  repeating 
coils  are  also  provided  at  both 

ends  of  the  local  loops  for  similar  Table  1 

reasons.  A  non-loaded  cable 
equalizer  is  provided  for  the  local 
cable  loops. 

Special  filters  are  required  for 
facilities  which  transmit  both 
program  and  carrier  frequencies, 
in  order  to  separate  the  two  fre- 
quency bands  at  the  line  termina- 
tions and  direct  them  into  then- 
proper  equipments.  Line  filters, 
designed  for  5000-cycle  program 
systems,  introduce  slight  delay 
distortion  in  the  program  fre- 
quency bands.  If  the  number  of 
line  sections  in  tandem  equipped 
with  these  filters  is  about  eight 
or  more,  a  delay  equalizer  for  each 
two  filters  is  necessary  to  main- 
tain the  time  of  propagation  of 
all  frequencies  in  the  program 
band  within  satisfactory  limits 
(not  to  exceed  about  0.3  milli- 
second difference  between  maxi- 


Classification  of  Facility 

Approximate 
Frequency 
Band  in  Cycles 

Intercity  Circuits 
1.  High  quality.  ..          

noo-  5,000 

2.  Medium  quality  

I    50-  8,000 
200-  3,500 

3    Speech,  only                             .        . 

300-  2  500 

Metropolitan  Area  Circuits 
1.  Studio-transmitter   circuits   for   AM 
stations 

50-  8  000  * 

2.  Studio-transmitter   circuits   for  PM 

stations 

50-15,000 

3.  Network  loops  —  between  studio  and 
point  of  connection  (toll  office)  with 

intercity  network  chanr^1!??    

f 

4.  Piek-up  circuits  —  between  points  of 
program  origin  and  studio  or  point  of 
connection  with  intercity  channels  .  .  . 

t 

*  Band  may  be  extended  to  higher  frequencies  if  specifically 
requested  by  the  customer. 

t  Equalized  at  request  of  the  customer  for  band  width  desired 
(usually  50-8,000  cycles). 


mum  and  minimum  delays  for  a  range  of  500  to  5000  cycles).  An  auxiliary  low-pass 
filter  may  be  provided,  if  required,  for  5000-cycle  systems,  to  effect  further  discrimination 
against  high-frequency  interference  from  the  line. 

An  auxiliary  low-pass  filter  may  be  used  on  8000-cycle  program  circuits  to  supplement 
the  low-pass  filter  of  the  carrier  line  filter  set  where  further  discrimination  is  needed. 

Noise  must  be  limited  in  all  program  circuits,  so  that  it  will  not  interfere  appreciably 
with  the  quality  of  the  broadcast. 

Pre  distorting  and  restoring  networks  are  employed  in  open-wire  program  systems, 
particularly  the  8000-cycle  system,  to  minimize  high-frequency  noise  from  nearby  carrier 
systems.  Predistortion  is  accomplished  by  introducing,  at  the  sending  end  of  the  cir- 
cuit, a  network  which  effectively  raises  in  volume  the  currents  above  1000  cycles  to  a 
higher  level  than  normal  for  line  transmission,  thus  increasing  the  signal-to-noise  ratio  at 
these  frequencies.  Since  the  power  at  the  higher  frequencies  is  relatively  small,  the 
amplifiers  are  not  overloaded  by  this  procedure.  The  restoring  network  at  the  receiving 
end  of  the  circuit  restores  the  predistorted  currents  to  their  original  amplitude  and  phase 
relation.  The  net  reduction  in  high-frequency  interference  is  equal  to  the  relative  losses 
introduced  by  the  restoring  network  at  the  frequencies  restored.  Figure  2  shows  the  char- 
acteristics of  these  networks,  which  are  of  the  lattice  type  and  are  composed  of  combina- 
tions of  inductance,  capacitance,  and  resistance  elements. 

Crosstalk  also  must  be  controlled.  Methods  of  limiting  crosstalk  include  such  items  as 
proper  selection  of  facilities,  maintenance  of  proper  levels,  and  avoidance  of  the  adverse 
effects  of  circuit  irregularities  which  may  develop  from  time  to  time. 

PROGRAM  FACILITIES  may  be  of  open  wire  or  cable,  assigned  for  program  use^in 
the  program  frequency  band,  or  such  facilities  may  consist  of  single-sideband  transmission 
over  cable  carrier  systems,  using  three  channels  of  a  twelve-channel  unit,  or  over  other 
carrier  system  channels. 

Open-wire  facilities  for  long-haul  broadcasting  usually  consist  of  165-mil  hard-drawn 
copper-wire  circuits,  although  128-  or  104-mil  wire  is  frequently  used,  where  available  and 


17-104 


TELEPHONY 


appropriate,  but  generally  for  the  shorter-haul  circuits.  Preference  is  given  to  8-in.  non- 
phantomed  pairs. 

Amplifiers  are  spaced  about  twice  as  frequently  (50  to  150  miles)  for  104  circuits  as  for 
165  circuits.  Open  wire,  being  more  subject  to  noise  and  crosstalk  than  cable  or  carrier, 
must  be  carefully  selected  and  maintained  in  order  to  provide  facilities  of  suitable 
quality. 

Cable  pairs  (non-quadded)  may  be  employed  for  long-haul  program  service.  The  latest 
type  of  cable  system  for  this  purpose  is  the  16-gage  B-22  loaded  system,  which  is  satis- 
factory for  a  band  of  35  to  8000  cycles. 

The  16-gage  B-22  system  operates  over  one-way  transmission  paths.  The  non-quadded 
cable  pairs  are  loaded  at  a  3000-ft  nominal  spacing  with  22-millihenry  coils.  The  attenu- 
ation-frequency characteristics  of  this  facility  are  shown  in  Fig.  19,  article  16,  and  the 
impedance,  cutoff,  and  velocity  values  are  shown  in  Fig.  18,  article  16.  Amplifiers  and 


ia 

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FREQUENCY  IN  KILOCYCLES  PER  SECOND 

FIG.  2.    Loss-frequency  Characteristics  of  Predistorting  and  Restoring  Networks — Open-wire  Program 

System  (Courtesy  Bell  System) 

associated  apparatus  are  located  at  about  50-mile  intervals.  Attenuation  and  delay 
equalizers,  as  required,  are  associated  with  each  line  amplifier  input  circuit.  A  supple- 
mentary adjustable  equalizer  is  usually  required  in  each  repeater  regulator  section,  be- 
cause of  the  variations  in  capacitance  and  conductance  of  the  different  cable  sections 
through  which  the  program  circuit  passes.  The  application  of  this  system  for  new  instal- 
lations may  be  limited  somewhat  in  the  future  by  the  newer  developments  in  single- 
sideband  carrier  program  transmission. 

High-quality  program  circuits  may  be  operated  over  cable  carrier  systems,  using  single- 
sideband  transmission  and  three  channels  of  the  system.  The  single-sideband  system  is 
designed  to  operate  over  as  many  as  ten  carrier  links  in  tandem.  Figure  3  shows  a  sche- 
matic of  a  single-sideband  program  terminal  arranged  for  transmitting  over  cable  carrier. 

Medium-quality  program  circuits  may  be  assigned  to  19-  or  16-gage  H-44  or  B-88  cable 
side  circuits,  but  non-linear  and  delay  distortion  in  the  repeaters  and  loaded  pairs  and 
transmission  variations,  particularly  at  high  and  low  frequencies,  limit  the  length  of  these 
facilities  to  about  300  miles  for  H-44  and  B-88  loaded  facilities.  Channels  of  the  cable 
carrier  system  may  be  used  for  medium-quality  service  if  not  more  than  one  link  of  these 
carrier  systems  is  used.  Non-loaded  cable  is  limited  to  relatively  short  lengths. 

METROPOLITAN-AREA  PROGRAM  FACILITIES,  as  indicated  in  Table  1,  are 
designed  to  provide  (1)  studio  to  transmitter  circuits  for  both  a-m  (amplitude-modulation) 
and  f-m  (frequency-modulation)  broadcasting  stations,  (2)  network  loops  between  the 
studio  and  the  point  of  connection  (usually  the  telephone  company  toll  office)  with  inter- 
city network  channels,  and  (3)  pick-up  circuits  between  the  point  of  program  origin  and 
the  studio  or  the  point  of  connection  with  intercity  network  channels. 

The  types  of  line  facilities  used  may  consist  of  non-loaded  exchange  or  toll  cable  pairs 
of  various  available  gages,  loaded  exchange  or  toll  cable  pairs  (where  loading  is  required 
for  transmission  reasons),  or  open-wire  pairs.  Metropolitan-area  program  circuits,  to  a 


PEOGEAM  SERVICE 


17-105 


large  extent,  employ  non-loaded  exchange  cable  pairs, 
pairs,   with   or  without  intermediate  ampli- 
fiers. 

Equalization  may  or  may  not  be  required, 
depending  on  the  band  width  to  be  trans- 
mitted, length  and  gage  of  facilities  involved, 
and  the  transmission  deviation  permissible 
over  the  frequency  band. 

For  studio  to  transmitter  circuits,  serving 
a-m  stations,  a  relatively  flat  transmission- 
frequency  characteristic  may  be  obtained 
between  50  and  8000  cycles,  using  standard 
equalizer  equipment,  for  sections  of  non-loaded 
cable  not  exceeding  about  21.5  miles  of  16-, 
10.0  miles  of  19-,  6.5  miles  of  22-,  5.0  miles  of 
24-,  or  4.2  miles  of  26-gage  cable  conductors. 
Longer  lengths  of  cable  may  be  divided  into 
sections  not  exceeding  the  above  lengths,  which 
are  then  treated  individually.  It  is  desirable, 
from  a  transmission  stand-point,  to  employ  a 
uniform  type  of  facility  for  program  circuits 
when  available. 

For  studio  to  transmitter  circuits,  serving 
f-m  stations,  standard  equalizer  arrangements 
provide  a  satisfactory  transmission  character- 
istic over  metropolitan-area  circuit  lengths 
usually  encountered.  In  one  instance,  such  a 
circuit,  consisting  of  24  miles  of  19-  and  22- 
gage  cable  pair  (non-loaded) ,  with  three  inter- 
mediate amplifiers,  was  equalized  to  limit  the 
overall  transmission  variation  to  1.9  db  for  a 
frequency  band  of  30  to  15,000  cycles.  Wider 
bands,  extending  to  18  or  20  kc,  have  been 
provided  in  special  cases  when  requested  by 
the  customer,  using  special  equalizing  methods. 

Carrier  loaded  (C  4.1  or  C  4.8  loading) 
cable  facilities  when  available  may  be  equal- 
ized for  metropolitan-area  circuits,  serving 
either  a-m  or  f-m  stations.  The  B  22  facilities 
are  suitable  for  frequencies  up  to  8000  cycles 
for  any  length  of  circuit  likely  to  be  encoun- 
tered, using  amplifiers  and  equalization  as  re- 
quired. 

SPECIAL  FEATURES  developed  for  pro- 
gram systems  include: 

(a)  Bridging  arrangements,  in  which  pro- 
vision is  made  for  connecting  one  or  more 
branch  program  circuits  to  the  main  program 
circuit  at  a  given  point. 

(6)  Monitoring,  in  which  attendants  may 
listen  in  and  supervise  programs  (to  insure 
satisfactory  operation)  at  designated  points  on 
the  broadcast  network. 

(c)  Reversals,  in  which  provision  is  made 
for  two-way  transmission  over  the  same  net- 
work facilities,  by  reversing  the  direction  of 
transmission  of  all  connected  one-way  ap- 
paratus at  will,  either  manually  or  automati- 
cally, and  under  control  of  the  telephone  com- 
pany or  the  customer. 

(£)  Order  wire  and  talking  arrangements, 
which  permit  the  various  control  points  and 
attendants  to  converse  readily  regarding  net- 
work operations  without  interfering  with  the 
broadcast  facilities. 


and  in  some   cases  open-wire 


17-106 


TELEPHONY 


SUBSCRIBER  STATIONS 


19.  SUBSTATION  EQUIPMENT 

SUBSCRIBER  STATION  EQUIPMENT  consists  of  a  large  number  of  types  and  designs 
of  subscriber  telephone  sets  and  auxiliary  apparatus,  essential  to  the  furnishing  of  a  com- 
plete telephone  service  in  the  most  economical  and  satisfactory  manner. 

The  basic  subscriber  telephone  set  is  assembled  in  a  number  of  designs  to  meet  the 
needs  of  different  services  and  the  subscriber's  convenience,  but  the  operating  principle  is 

primarily  the  same  for  all  these  sets.  The 
basic  set  consists  principally  of  a  trans- 
mitter, receiver,  induction  coil,  condens- 
ers, ringer,  and  spring  assembly  (switch- 
hook),  suitably  mounted  in  a  metal  or 
plastic  housing.  The  set  will  also  have 
a  dial  if  it  is  connected  to  a  dial  exchange 
or  unit,  and  for  magneto  telephones  a 
dry-cell  battery  for  transmitter  current 
and  a  magneto  generator  for  signaling 
must  be  provided. 

THE  TELEPHONE  TRANSMITTER 
is  designed  to  receive  airborne  sound 
waves  of  various  frequencies  and  convert 
them  into  electrical  waves  of  similar  fre- 
quencies for  transmittal  over  a  telephone 
circuit.  It  is  essentially  a  device  which 
makes  it  possible  for  relatively  weak 
sound  energy  to  control  electrical  energy 
of  greater  average  strength. 

The  modern  transmitter  unit  employs 
an  insulated,  spherical-shaped  carbon 
chamber  holding  the  carbon  granules,  and 
a  very  light  metal  conical  diaphragm,  with 
a  dome-shaped  center,  which  is  positioned 
in  the  carbon  chamber  in  such  a  way  as 
to  hold  the  granules  in  the  chamber.  The 
dome  and  chamber  are  the  front  and 
back  electrodes,  respectively,  of  the  unit. 
Figure  1  shows  a  cross-sectional  view  of 
the  Western  Electric  Co.  No.  Fl  unit. 
The  electrode  surface  area  in  contact  with  the  carbon  granules  is  the  same,  regardless  of 
the  position  in  which  the  transmitter  is  used.  The  No.  Fl  unit  has  a  resistance  (new) 
of  about  30  to  40  ohms  and  is  applicable  to  any  type  of  subscriber  telephone  set  of 
Western  Electric  Co.  make.  Transmitter  current  should  not  exceed  about  100  milliam- 
peres  for  normal  usage  and  life.  A  small  condenser  is  usually  connected  directly  across 


PAPER  BOOKS ~ 


DIAPHRAGM 


SILK  CLOSURE  AND 
CONTACT  MEMBER' 


MOVINS-FRONT 
ELECTRODE     ' 
ATTACHED  TO 
DIAPHRAGM 

OILED-SILK 
MEMBRANE 


DIAPHRAGM 


BRASS  GRID 


—  INSULATORS 


FIG.  1.     Cross-sectional  View  of  Modern  Non-posi- 
tional Transmitter  Unit  (Courtesy  Bell  System) 


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FREQUENCY   IN    KILOCYCLES   PER  SECOND 

FIG.  2.     Transmitter  Response  Characteristics  (Courtesy  Bell  System) 

the  transmitter  electrodes  of  the  Fl.unit,  when  used  in  handsets,  to  minimize  packing 
(welding  together)  of  carbon  granules  as  the  result  of  arcing  in  the  carbon  chamber  when 
the  transmitter  current  is  rapidly  interrupted  by  switchhook  flashing  or  other  operations. 


SUBSTATION  EQUIPMENT 


17-107 


AIR  CHAMBER 


WJNDJIvKS 


PERMALLOY 

POLE  FHECE 


The  response  of  all  makes  of  the  modern  non-positional  transmitter  is  greatly  improved 

°TT^e™T%S2eS'  as  ^own  for  Western  Electric  Co.  instruments  in  Fig.  2. 

IMJi,  l&iJiPHONE  RECEIVER  is  designed  to  receive  electrical  waves  of  various  fre- 
quencies and  convert  them  into  sound  waves  of  similar  frequencies  for  the  listener's  ear. 

It  is  essentially  a  reconverter  from  electric  to  sound  energy  (within  its  frequency  range) 

and  is  a  necessary  complement  of  the  transmitter  in  the  transmission  of  speech. 

The  modern  receiver  is  generally  of  the  unit  or  capsule  design,  which  is  applicable  to 

various  types  of  telephone  receivers  and 

assembled  sets  as  made  by  the  respec-  ifl^Kfffr        .xREMAUjor  BAR  MAGNET 

tive  manufacturers.     One  make  (W.E. 

Co.  HA1)  consists  principally  of  a  bipolar 

permanent  magnet,  with  the  parts  as- 
sembled in   a  zinc   alloy  frame.     The 

diaphragm  of  Permendur  is  seated  on  a 

ring  projection  of  the  frame,  just  above 

the  magnet,  with  an  air  chamber  of  def- 
inite volume  behind  it.    This  chamber 

has  a  small  outlet  hole  covered  with  a 

silk  disk  of  specified  acoustic  impedance. 

The  diaphragm  also  has  an  air  chamber 

of  definite  volume  between  it  and  the 

receiver  cap,  which  latter  has  six  holes 

of  definite  length  and  area.     The  un- 

clamped  diaphragm  thus  rests  between 

two  air  chambers  of  specified  volumes 

and  outlet  impedances,  and  variations 

in  receiver  efficiency  with  temperature 

changes  as  well  as  diaphragm  freezing  to 

the  pole  pieces  are  practically  eliminated. 
The  HA1  receiver  unit  has  a  working 

impedance  at  1000  cycles  of  about  140 

ohms  with  a  positive  angle  of  60°.     Figure  3  shows  a  cross-sectional  view  of  this  type  of 

receiver  unit,  and  Fig.  4  shows  its  response  characteristic  as  well  as  that  of  the  older  type 

W.E.  Co.  No.  557  receiver. 

The  HA1  receiver  is  affected  adversely  by  d-c  flow  of  either  polarity,  which  for  100 

milliamperes  amounts  to  about  4.5  and  6.0  db  loss  in  volume  efficiency  for  the  opposing  and 

aiding  directions,  respectively,  of  current  flow. 

The  anti-sidetone  type  induction  coil  is  now  generally  employed,  in  place  of  the  older 

sidetone  type.     Figure  5  shows  schematic  diagrams  of  both  sidetone  and  anti-sidetone 

station  circuits  and  the 
direction  of  instantane- 
ous current  flow  for  both 
the  transmitting  and  re- 
ceiving conditions- 

In  the  sidetone  connec- 
tion (Fig.  5,  circuit  1), 
the  speech  currents  pro- 
duced in  the  transmitter 
divide  between  the  A  and 
B  windings  (solid  arrows) . 
The  current  in  each  of 
these  windings  induces  a 
voltage  in  the  other  wind- 
ing (dashed  arrows) . 
The  two  currents  in  A 
combine  to  flow  out  over 
the  line,  and  the  two 
currents  in  B  combine  to 
flow  through  the  receiver 


ZIlsJC  ALLOY  FRAME 


PIG.  3.    Cross-sectional  View  of  Modern  Receiver  Unit 
(Courtesy  Bell  System) 


RESPONSE  IN  DECIBELS 
(0  DECIBELS  =  1  BAR  PER  WATT) 

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FREQUENCY  IN  KILOCYCLES  PER  SECOND 

PIG.  4.'    Receiver  Response  Characteristics   (Courtesy  Bell  System) 


and  produce  sidetone.  When  receiving,  the  incoming  current  divides  between  the  trans- 
mitter (T)  and  the  receiver  C5)  branches  (solid  arrows).  The  current  in  A  induces  a 
voltage  in  B,  which  establishes  an  opposing  current  in  B  (dashed  arrow),  but  the  incom- 
ing line  current  in  B  is  larger,  so  that  the  resultant  current  through  R  is  sufficient  to 
actuate  the  receiver  diaphragm. 

The  sidetone  reduction  connection  of  the  sidetone  coil  employed  on  short  loops  having 


17-108 


TELEPHONY 


the  sidetone  type  of  subscriber  set  is  shown  in  Fig.  5,  circuit  2.  With  this  connection  the 
transmitter  current  cannot  flow  directly  through  the  receiver  (R)  branch,  but  some  in- 
duced current  does  flow  through  R  because  of  the  inductive  coupling  between  windings  A 
and  B,  Since  transmitter  current  does  not  flow  directly  through  B,  this  winding  does  not 
induce  a  voltage  in  A,  such  as  occurred  in  the  sidetone  connection,  and  the  transmitter 
line  current  is  not  thus  aided,  resulting  in  lower  transmitting  efficiency.  In  receiving, 
the  incoming  line  current  does  not  enter  the  B  winding  directly  but  flows  through  winding 
A,  establishing  an  induced  voltage  from  A  to  5,  which  causes  received  current  to  flow 
through  R.  Thus,  the  induced  R  current  is  not  opposed  by  line  current  flowing  directly, 
as  was  the  case  in  the  sidetone  connection,  resulting  in  slightly  higher  receiving  efficiency. 


SIDETONE  CIRCUITS 


-STANDARD   CONNECTION 


SIDETONE  REDUCTION 
CONNECTION 


RECEIVING  (CIRCUIT  3) 


TRANSMITTING  (CIRCUIT  3) 


CIRCUIT   3 
FIG.  5.     Sidetone  and  Anti-sidetone  Station  Circuits  (Courtesy  Bell  System) 

In  the  anti-sidetone  connection  (Fig.  5,  circuit  3),  a  third  winding  of  relatively  high  re- 
sistance, so  that  the  winding  has  both  an  inductance  (C)  and  resistance  (N)  component, 
has  been  added  to  the  sidetone  circuit. 

In  receiving,  the  incoming  current  divides  between  the  transmitter  (T)  branch  and 
winding  B  (solid  arrows).  The  current  in  B  divides  between  the  receiver  (R}  branch 
and  the  winding  C  (solid  arrows).  The  currents  in  A  and  B  induce  voltages  in  C,  and, 
by  properly  proportioning  the  coil  winding  relations,  the  current  in  C  (dashed  arrow) 
resulting  from  these  combined  induced  voltages  is  opposite  and  about  equal  to  the  current 
flowing  directly  in  C.  The  C  winding  thus  has  no  appreciable  effect  on  receiving  volume, 
and  the  receiving  efficiency  of  both  the  sidetone  and  anti-sidetone  circuits  is  about  the 
same.  However,  owing  to  sidetone  suppression,  the  effective  receiving  losses  for  the  anti- 
sidetone  circuit  are  substantially  less. 

In  transmitting,  the  transmitter  (T)  output  current  divides  between  the  windings  A 
and  B,  and  the  current  entering  B  divides  between  the  receiver  (R)  branch  and  the  C 
winding  (solid  arrows).  These  currents  in  A  and  B  induce  voltages  in  B  and  A,  respec- 
tively, resulting  in  an  induced  current  flowing  in  an  aiding  direction  in  the  line  and  in 
B  (dashed  arrows) .  The  induced  current  in  B  divides  between  R  and  winding  C  (short 
dashed  arrow) .  Also,  there  is  induced  in  C  a  voltage  (resulting  from  the  currents  flowing 
in  A  and  B  directly  from  the  transmitter)  which  establishes  a  current  in  C  (long  dashed 
arrow) .  The  combined  currents  in  C  flow  through  R  in  a  direction  opposite  to  the  current 
(solid  arrow)  which  comes  directly  from  the  transmitter.  Since  these  opposing  currents 
are,  by  design,  about  equal,  the  receiver  is  not  appreciably  affected  during  transmitting, 
and  sidetone  is  practically  eliminated.  The  transmitting  volume  efficiency  of  both  the 
anti-sidetone  and  sidetone  circuits  is  about  the  same,  but  the  effective  transmitting  losses 
for  the  anti-sidetone  set  are  substantially  less,  principally  because,  when  less  sidetone  is 
heard  by  the  talker,  he  unconsciously  raises  his  voice  level  until  the  sidetone  heard  is 
about  equal  to  what  he  is  accustomed  to  in  ordinary  conversation.  Increased  voice  level 
results  in  transmission  gain.  , 


SUBSTATION  EQUIPMENT  17-109 

The  range  of  line  impedance  conditions  is  usually  much  greater  for  the  local  battery 
anti-sidetone  set  which  is  used  on  the  longer  loops  into  outlying  and  rural  areas.  In  order 
to  provide  an  approximate  balance  for  these  line  conditions,  the  local  battery  anti-side- 
tone  circuit  is  designed  with  a  balancing  network  that  can  be  connected  for  a  line  im- 
pedance of  600,  900,  or  1500  ohms  (angle  50°). 

Under  average  conditions,  and  depending  upon  the  amount  of  sidetone  reduction  which 
may  be  effected  in  any  particular  case  and  upon  the  circuit  design,  a  transmission  im- 
provement of  about  5  db  may  be  expected  from  use  of  the  anti-sidetone  in  place  of  the 
sidetone  circuit. 

THE  RINGING  CIRCUIT  of  a  common-battery  subscriber  set  includes  a  ringer  and 
condenser,  but  a  condenser  may  or  may  not  be  required  in  this  circuit  for  a  magneto  set. 
The  common-battery  sidetone  circuit  has  a  single  condenser  for  use  in  both  the  receiver 
and  ringer  circuit,  but  with  development  of  the  anti-sidetone  circuit  a  separate  condenser 
is  used  hi  the  receiver  and  ringer  circuits.  "With  the  two-  (split-)  condenser  arrangement 
and  high-impedance  ringers,  the  susceptiveness  of  the  subscriber  set  to  incoming  line  noise 
is  materially  reduced  over  the  single-condenser  set,  especially  on  balanced  party-lines  (same 
number  and  type  of  ringers  connected  to  ground  on  each  side  of  the  line).  Ringers  are 
manufactured  in  a  large  number  of  types  (usually  with  permanent  magnet  yokes  and  bias- 
ing springs)  and  impedances  for  use  with  the  several  signaling  systems  and  to  meet  signal- 
ing requirements.  The  types  of  signaling  systems  used  include  straight  line  (20  or  16  2/3 
cycle),  harmonic,  superimposed,  and  pulsating  ringing,  of  which  straight  line  ringing  is 
commonly  used  for  individual  line,  two-party  selective,  four-party  semi-selective,  divided 
code,  or  non-selective  bridged  stations. 

Harmonic  ringers  are  designed  for  frequencies  of  16.6,  20,  25,  30,  33.3,  42,  50,  54,  60, 
66,  and  66.6  cycles.  These  ringers  have  reeds  of  different  weights  which  respond  only  to 
the  frequency  for  which  they  are  designed.  The  harmonic  system  provides  selective 
signaling  for  up  to  five  bridged  or  ten  grounded  ringing  stations  (five  connected  to  ground 
on  each  side  of  the  line) . 

Superimposed  ringing  requires  the  superimposing  of  d-c  potentials  (positive  and  nega- 
tive) on  the  regular  a-c  ringing  current  in  order  to  raise  the  peak  positive  and  negative 
voltages  sufficiently  to  break  down  a  three-element,  cold-cathode,  vacuum-tube  gap, 
which  is  connected  from  line  to  ground  at  each  station.  The  tube  may  be  so  connected 
that  its  gap  will  break  down  when  either  the  positive  or  negative  combined  potential  is 
applied,  but  not  both.  The  ringer  is  connected  between  one  element  of  the  tube  and 
ground  so  that,  when  the  tube  functions,  the  a-c  ringing  current  will  operate  the  ringer. 
This  type  of  ringing  is  used  in  four-party  selective  and  eight-party  semi-selective  service. 
The  control  gap  of  one  type  of  tube  has  a  nominal  breakdown  of  70  volts,  and  the  nominal 
main  (ringing)  gap  sustaining  voltage  is  75  volts. 

A-c  relays,  first  used  in  this  type  of  service,  generally  have  been  replaced  by  the 
tube. 

Pulsating  ringing  employs  positive  and  negative  pulsations  sent  out  over  the  line  from 
the  office.  The  station  ringers  are  biased  with  a  spring  so  that  they  operate  when  either 
the  positive  or  negative  pulsations  are  applied,  but  not  both.  This  type  of  ringing  may 
be  used  for  four-party  selective  or  eight-party  semi-selective  service. 

Ringer  impedances  are  designed  to  meet  various  service  conditions.  The  low-impedance 
group  will  have  d-c  resistances  of  from  1000  to  about  2500  ohms,  and  the  high-impedance 
group  of  from  3500  to  about  5700  ohms  or  more. 

Usually  each  ringer  has  a  biasing  spring,  one  end  of  which  is  attached  to  the  armature 
and  the  other  end,  by  means  of  a  winding  cord,  to  an  adjusting  stud  on  the  ringer  frame. 
Thus,  the  armature  can  be  tensioned  or  biased  to  the  degree  necessary  for  the  type  of 
service  involved,  and  to  avoid  cross  rings  and  also  bell  taps  during  switchhook  and  dial 
operation.  Ringers  may  be  of  the  polarized  or  non-polarized  type,  depending  upon  the 
type  of  ringing  system  employed. 

The  number  of  ringing  bridges  across  a  line  or  from  either  side  of  a  line  to  ground  is 
limited  to  a  total  capacitance  of  the  ringer  condensers  not  to  exceed  2  /if  where  not  more 
than  eight  such  bridges  are  involved  on  non-polarized  lines. 

Condensers  are  usually  of  2-/if  capacitance  when  used  in  single-condenser  sets  and  in 
the  transmission  circuit  of  two-condenser  sets.  The  ringer  circuit  of  these  latter  sets  may 
have  a  condenser  of  0.5-  to  1.0-^f  capacitance,  depending  upon  the  ringer  impedance,  or 
for  some  types  of  magneto  stations  the  condenser  may  be  omitted. 

Hand  generators  of  the  three-  or  five-bar  type,  depending  upon  the  length  of  line  and 
number  of  bridged  stations,  are  generally  necessary  in  magneto  subscriber  sets  for  signal- 
ing the  office  or  another  subscriber  on  the  same  line.  These  generators  carry  a  spring 
assembly  which  functions  to  open  and  close  the  generator  circuit  across  the  line  when  the 


17-110 


TELEPHONY 


generator  is  idle  and  operating,  respectively.    Figure  6  shows  a  view  of  a  five-bar  gen- 
erator (Stromberg-Carlson  make)  . 

The  switchhook  spring  assembly  in  subscriber  sets  generally  consists  of  two  pairs  of 
make-break  springs  which  automatically  close  and  open  the  transmitter  and  receiver 
circuits  when  the  handset  is  lifted  from  and  restored  to  its  cradle.  These  springs  are  de- 
signed to  retain  their  adjustments  over  long  periods  of  time.  One  manufacturer  is  using 

heavy-gage,  phosphor-bronze  with  precious- 
metal  contacts,  which  provide  positive,  non- 
microphonic  contact. 

Dials  are  required  at  all  dial  stations 
served  by  mechanical  offices  or  private 
automatic  exchanges  of  any  type.  The 
dial  is  the  subscriber's  only  means  at  dial 
stations  of  securing  connection  with  other 
dial  subscribers  or  with  toll  or  assistance 
operators  in  his  exchange  area.  Figure  7 
shows  front  and  rear  views  of  one  type  of 
dial  assembly.  The  front  view  shows  the 
finger  wheel  and  stop  and  the  number  plate 
with  its  letter  and  number  designations 
appearing  through  the  finger  wheel  holes. 
The  rear  view  shows  the  spring  assembly, 
consisting  of  shunt  (off  normal)  springs 
which  short-circuit  the  transmitter  and 
open  the  Deceiver  circuits  during  dialing 
and  the  impulse  springs  which  open  and 
close  the  subscriber  line  circuit,  thus  causing 
pulses  of  current  to  flow  in  this  circuit  at  an  average  rate  of  about  10  pulses  per  second. 
The  finger  wheel,  having  been  turned  to  the  finger  stop  for  any  selected  letter  or  number, 
is  released,  and  as  it  returns  to  normal  under  spring  action  it  is  geared  to  operate  the  pulse 
pawl  at  a  speed  controlled  by  the  governor.  The  pulse  pawl  alternately  opens  and  per- 
mits the  impulse  springs  to  close  as  many  times  as  there  are  units  in  the  digit  pulled. 

Small  filters  consisting  of  inductance,  capacitance,  and  resistance  are  generally  bridged 
across  the  impulse  spring  contacts  in  order  to  reduce  dialing  interference  to  nearby  radio 
receivers. 

Housings  of  various  types  are  provided,  in  which  the  required  subscriber  telephone  set 
units  described  above  are  assembled.  These  housings  are  made  of  zinc  alloy,  plastics, 
steel  for  baseplates,  and  other  materials,  including  rubber  for  cushioning.  The  various 
units  are  usually  assembled  on  a  universal  baseplate,  suitable  for  wall  or  desk-type  sets. 


berg-Carlson  Co.) 


Rear  of  Dial 


Case, 


ulse  Contacts 
'ulse  Pawl 


inger  Stop 
'Finger  Wheel 


Off  Normal 
Contacts 


FIG.  7.    Front  and  Rear  View  of  Dial  as  Used  with  Dial  Type  Subscriber  Telephone  Set  (Courtesy  Bell 

System) 

In  one  type  of  set,  made  by  the  Kellogg  Switchboard  and  Supply  Co.,  plug-in  type  in- 
duction coil  (AST)  and  condenser  units  permit  ready  replacements  to  be  made  by  pulling 
out  the  unit  to  be  replaced  and  plugging  in  the  new  unit.  The  design  of  this  company's 
set  includes  a  universal  permanently  wired  circuit,  consisting  of  a  stamped  metal  grid 
nested  in  the  underside  of  an  interconnecting  block  which  mounts  on  a  baseplate.  This 
grid  is  connected  through  the  block  to  screw  terminals  and  pin  jacks  mounted  on  the 
upper  side  of  this  block.  The  induction  coil  has  a  three-way  and  the  condenser  a  two-way 
switching  unit  to  permit  convenient  circuit  adjustments. 


SUBSTATION  EQUIPMENT 


17-111 


Some  companies  are  now  providing  non-interfering  or  press-to-talk  features  in  telephone 
sets  intended  primarily  for  use  on  multiparty  rural  lines,  to  permit  listening  on  the  line 
without  the  transmitter  being  cut  into  the  circuit  and  without  interfering  with  the  ringing 
and  dialing  or  with  conversations  in  progress. 


HAND  SET  CORD 


*    IF  FILTER  IS  NOT  USED  CONNECT  THIS  LEAD  TO  THE 
*Y*  TERMINAL  OF  THE  DIAL 

FIG.  8.    Schematic  of  Wiring  of  Subscriber  Hand  Telephone  Set — Common  Battery  Talking — Common 
Battery  Signaling  (Courtesy  BeU  System) 

Figures  8  and  9  show,  respectively,  schematics  of  the  wiring  of  a  typical  common  bat- 
tery talking-common  battery  signaling  (CBT-CBS)  and  a  typical  local  battery  talking- 
common  battery  signaling  (LBT-CBS)  subscriber  hand  telephone  set  for  individual,  two- 
party  selective,  or  four-party  semi-selective  service.  The  dial  and  filter  circuits  are  re- 
quired for  dial  service  only. 


*1F  FILTER  IS  NOT  USED 
CONNECT  THIS  LEAD  TO  THE 
*Y*  TERMINAL  OF  THE  DIAL 

FIG.  9.    Schematic  of  Wiring  of  Subscriber  Hand  Telephone  Set — Local  Battery  Talking — Common 
Battery  Signaling  (Courtesy  Bell  System) 

SUBSCRIBER  SERVICES  involving  other  than  the  regular  subscriber  telephone  set 
just  described  require  a  wide  variety  of  substation  apparatus.  Owing  to  limited  space  in 
this  handbook,  only  brief  mention  can  be  made  of  some  of  the  principal  services  currently 
rendered,  which  are: 

1.  Station  wiring  plans,  whereby  individual  central  office,  PBX,  or  private  lines  may 
be  connected  at  one  or  more  telephone  stations  of  which  one  or  more  or  none  may  have  a 


17-112  TELEPHONY 

regular  attendant.  In  these  cases,  each  station  telephone  may  be  equipped  in  its  base 
with  several  pushbutton  type  keys,  or  separate  key  boxes  may  be  provided  which  can  be 
used  for  originating,  answering,  intercepting,  holding,  or  transferring  central-office  calls 
on  one  or  more  lines,  or  connections  can  be  established  between  stations  without  using 
either  a  central-office  or  PBX  line.  Various  arrangements  for  controlling,  locking  out, 
grouping,  or  extending  station  calls  are  also  available.  Signaling  between  stations  is 
usually  by  a  pushbutton  and  buzzer  circuit  with  battery  or  low  voltage  alternating  cur- 
rent derived  from  the  commercial  power  supply.  Operating  power  may  be  supplied  lo- 
cally or  from  the  central  office. 

2.  Loudspeaking  and  distant  talking  systems  may  be  employed  between  a  master  and  one 
or  more  regular  stations,  facilitating  communication,  and  such  systems  may  be  associated 
with  regular  telephone  facilities  and  equipments.     These  systems  require  amplifiers, 
which  are  associated  with  each  loudspeaking  and  distant  talking  telephone  set,  to  provide 
adequate  volume  from  the  loudspeaker.     Commercial  power  is  usually  employed  for  this 
equipment. 

3.  An  operator-type  transmitter  and  receiver  set,  with  an  associated  key  and  jacks,  may 
be  provided  at  subscriber  stations  to  facilitate  handling  messages  over  the  telephone. 

4.  Loudspeaker  conference  service  may  be  provided,  where  groups  of  people  desire  to 
listen  to  a  local  or  distant  talk  and  are  assembled  at  one  or  more  points.     An  amplifier 
and  loudspeaker  with  a  suitable  subscriber  set  and  switching  keys  are  required  for  this 
service.     Commercial  power  is  used  for  this  equipment. 

5.  Coin  collectors  of  various  types  are  located  in  public  places  where  customers  may 
place  local  or  long-distances  calls,  either  by  prepayment  or  postpayment  of  the  charges. 
Some  of  these  collectors  are  placed  in  booths;  others,  in  the  open  at  locations  convenient 
for  public  use. 

6.  Subscriber  amplifier  deaf  set  equipment  provides  for  amplification  of  the  incoming 
voice  currents  before  reaching  the  subscriber's  receiver.     The  incoming  volume  level  is 
raised  sufficiently,  in  many  cases  of  deafness,  for  the  subscriber  to  carry  on  a  telephone 
conversation  which  would  otherwise  not  be  possible. 

7.  Code  calling  systems  sound  code  signals  at  various  points  throughout  a  subscriber's 
establishment  to  notify  certain  employees  or  officials  that  they  are  wanted  at  the  telephone 
or  for  some  other  reason.     The  controlling  station  on  this  system  is  usually  located  at  the 
subscriber's  PBX  switchboard  and  is  operated  by  the  PBX  attendant  as  a  separate  sys- 
tem, or  the  system  may  be  actuated  directly  from  a  dial  PBX  or  PAX  system  without  an 
attendant. 

This  system  requires  code  sending  and  station  signal  equipment,  properly  located  and 
interconnected  by  wiring.     Commercial  a~c  power  supply  is  employed  for  operating  it. 

8.  Loudspeaker  paging  systems  are  designed  to  provide  a  means   of  simultaneously 
transmitting  messages  or  announcements  verbally  from  a  central  location  to  a  number  of 
points  within  an  establishment.     These  systems  have  a  variety  of  uses  from  summoning 
a  person  to  a  telephone  or  directing  employee  activities,  to  providing  information  to  a 
limited  area. 

9.  Subscriber  telephone  sets  for  explosive  atmospheres  are  available  for  use  in  mines,  oil 
refineries,  or  munition  plants.     The  equipment  is  designed  to  prevent  sparking  of  the 
various  parts  under  operation,  causing  explosions. 

10.  Outdoor-type  telephone  sets  are  provided  for  mounting  in  outdoor  places  for  fire, 
police,  taxicab,  and  other  services.     These  sets  are  enclosed  in  cast-iron  or  wooden  hous- 
ings, which  protect  the  equipment  from  weather  conditions. 

11.  Sound  powered  telephones  are  provided  in  locations  where  it  is  desired  to  avoid  pos- 
sible central  energy  failures  and  where  the  distances  between  stations  are  relatively  short, 
such  as  in  an  establishment,  on  a  ship,  or  as  portable  field  telephone  equipment  where 
batteries  are  not  desirable.     These  telephones  operate  through  action  of  sound  waves 
striking  the  transmitter  diaphragm  and  causing  a  variation  in  a  magnetic  field  which  pro- 
duces electric  currents  in  the  telephone  circuit  of  frequencies  similar  to  those  in  the  sound 
waves.     The  receiving  apparatus  functions  quite  like  the  regular  telephone  receiver. 

12.  Program  distributing  systems  are  designed  to  furnish  program  material,  either  from 
broadcasting  stations  or  central  program  points,  to  hospitals,  schools,  hotels,  business 
establishments,  factories,  homes,  and  many  other  locations.     Program  material  may  con- 
sist of  music,  speeches,  announcements,  and  various  other  features  of  interest.     This 
material  is  usually  transmitted  over  wire  lines  from  its  source  to  one  or  more  common 
amplifiers  and  thence  distributed  to  subscribers  over  wire  lines  to  loudspeakers  at  the 
various  locations. 


SUBSCRIBER  STATION   PROTECTION  17-113 

20.  SUBSCRIBER  STATION  PROTECTION 

Substation  protection  is  required  to  protect  subscribers  and  substation  equipment  from 
dangerous  voltages  and  currents.  The  telephone  plant  is  designed  to  withstand,  with 
some  margin,  its  normal  operating  currents  and  voltages.  But  there  are  other  sources  of 
electrical  power,  chiefly  lightning  and  commercial  power  linea,  which  may  under  certain 
conditions  impress  large  and  destructive  voltages,  with  resulting  excessive  currents,  on  the 
telephone  plant,  either  by  direct  contact  or  by  induction. 

Danger  from  lightning  in  cities  is  less  than  in  sparsely  settled  areas  because  of  the  shield- 
ing effect  of  buildings,  trees,  and  various  overhead  structures.  The  danger  from  power- 
line  ^contacts  with  aerial  telephone  plant  is  always  present  where  the  two  types  of  plant 
are  in  close  proximity,  although  material  progress  has  been  made  in  lessening  this  hazard 
over  the  years  by  improved  methods  of  construction  and  protection.  Stations  connected 
to  lines  which  are  exposed  to  more  than  250  volts  between  wires  usually  require  protection 
and  are  classed  as  exposed  stations. 

The  maximum  voltages  most  commonly  impressed  on  subscriber  lines  for  telephone 
purposes  range  from  24  to  50  volts  direct  current  and  from  75  to  175  volts  alternating 
current.  Direct-current  flow  over  the  subscriber  loop  (mdividual  line  service)  does  not 
usually  exceed  150  to  200  milliamperes  and  in  most  cases  is  less  than  about  100  milli- 
amperes. 

One  type  of  substation  'protector  widely  used  consists  principally  of  two  pairs  of  carbon 
protector  blocks  (open  space  cutouts)  and  two  line  fuses  (one  pair  of  blocks  and  one  fuse 
for  each  side  of  the  line) ,  with  associated  spring  and  terminal  holders,  mounted  on  a  por- 
celain base.  Another  design  employs  two  metal  plates,  each  of  which  connects  with  one 
side  of  the  Hne  and  has  a  sawtooth  inner  edge  positioned  0.004  in.  from  a  grounded  carbon 
block.  The  discharges  take  place  across  this  gap.  The  ground  electrode  of  the  protector 
must  be  well  grounded  in  every  case. 

CARBON"  PROTECTOR  BLOCKS  (sometimes  designated  as  lightning  arresters  or 
dischargers)  are  manufactured  hi  several  designs,  with  the  gap  between  the  Hne  and  ground 
block  varying  from  about  0.003  to  0.075  in.  or  more.  The  dielectric  between  the  blocks 
may  be  air,  or  mica  or  acetate  separators  may  be  used.  One  design  common  throughout 
the  United  States  consists  of  a  grooved  porcelain  block  with  carbon  insert  for  the  line 
contact,  and  a  solid  flat  (plain)  carbon  block  for  the  ground  contact.  The  Hne  spring 
bears  against  the  carbon  insert  set  in  the  porcelain  frame,  and  the  porcelain  frame  bears 
against  the  ground  block.  The  carbon  insert  is  held  in  place  by  a  glass  cement  of  low 
melting  point,  and  is  forced  against  the  ground  block  when  arcing  across  the  gap  is 
sufficient  to  soften  the  cement. 

The  carbon  insert  is  accurately  positioned  in  its  porcelain  frame  to  provide  a  0.003-in. 
gap  for  substation  protection,  or,  when  used  at  the  junction  of  cable  and  open  wire  lines, 
this  gap  is  0.006  in.  The  peak  breakdown  voltages  for  these  gaps  are  about  350  average 
and  550  maximum  for  the  0.003-in.  gap  and  about  710  average  and  1080  maximum  for  the 
0.006-in.  gap. 

The  fuses,  being  connected  in  series  with  the  line  and  on  the  line  side  of  the  protector 
blocks,  are  designed  to  open  each  side  of  the  line  when  the  protector  blocks  discharge  heav- 
ily or  break  down  completely  and  when  the  resulting  current  through  the  fuses  exceeds 
their  ratings,  which  are  usually  5  or  7  amperes,  although  fuses  of  other  amperage  may  be 
used,  depending  upon  operating  company  requirements. 

Grounding  of  the  protector  requires  a  reasonably  low-resistance  ground  (less  than  about 
25  ohms),  which  may  be  obtained  at  the  subscriber's  premises  by  connecting  to  (1)  the 
public  water  pipe  system,  (2)  a  private  water  pipe  or  well  casing  system,  or  (3)  a  driven 
ground  (ground  rod  or  pipe) ,  preference  being  given  in  the  order  named. 

A  common  ground  with  the  secondary  neutral  of  power  distribution  wiring  is  usually 
employed  at  the  subscriber  premises  unless  adequate  separation  between  the  telephone 
and  power  wiring  on  the  premises  can  be  maintained.  This  is  to  avoid  excessive  poten- 
tials being  impressed  on  the  telephone  wiring,  where  separate  grounds  are  employed, 
should  there  develop  abnormal  currents  in  the  secondary  neutral  due  to  its  becoming 
crossed  with  the  primary  power  circuit. 

Where  extensive  public  water  pipe  or  other  systems  having  a  low  resistance  to  ground 
are  available,  common  grounding  should  be  used.  Where  such  systems  are  not  available, 
consideration  must  be  given,  in  deciding  on  a  separate  or  common  ground,  to  the  prob- 
ability of  high  potentials  being  impressed  on  the  telephone  wiring  in  case  of  high  currents 
in  the  secondary  neutral. 

Figure  10  shows  one  type  of  substation  protector,  widely  used,  with  a  fuse  mounted 
along  each  side  and  the  protector  blocks  held  in  a  spring  assembly  set  in  a  well  in  the 


17-114 


TELEPHONY 


center  of  the  porcelain  mounting  block.     The  metal  cap  shown  screws  down  over  the 
blocks,  excluding  dirt  and  moisture. 

Substations  located  at  power  stations  generally  require  special  protection.  Whenever 
an  abnormal  condition  on  a  power  system  results  in  ground  current  between  the  power 
station  and  an  outside  point  on  the  system,  the  ground-potential  rise  at  the  power  station 
above  that  of  distant  grounds  will  depend  on  the  IZ  drop  in  the  power-station  ground. 


FIG.  10.     Substation  Protector  (Courtesy  BeU  System) 

Such  a  potential  rise  might  exceed  the  breakdown  potential  of  the  usual  telephone  pro- 
tector and  render  the  telephone  circuit  inoperative  at  a  time  when  it  is  most  needed.  Spe- 
cial protective  devices,  consisting  of  a  neutralizing  transformer  or  remote  grounding  of  the 
telephone  protector,  may  be  considered  in  determining  the  type  of  special  protection  to 
use. 

21.  PRIVATE  BRANCH  EXCHANGE  EQUIPMENT 

Private  branch  exchange  (PBX)  equipment,  as  discussed  here,  includes  manual  and  dial, 
cordless  and  cord,  attended  and  unattended  switchboards,  whatever  particular  designa- 
tions (such  as  PAX  or  PABX)  are  given  to  them  by  various  companies.  These  boards 
range  in  size  from  the  small  ten-line  system  to  the  largest  multiple  type  having  a  capacity 
of  about  3000  lines  or  more. 

MANUALLY  OPERATED  PBX  BOARDS  may  be  of  the  cordless  type  serving  a  few 
stations  or  of  the  cord  type  for  the  larger  installations.  These  boards  may  also  be  oper- 
ated in  conjunction  with  PBX  dial  equipment. 

A  typical  type  of  manual  cordless  common-battery  PBX  board  has  a  capacity  of  twelve 
extensions  (stations),  five  central-office  trunks,  and  five  connecting  circuits.  Each  ex- 
tension and  each  trunk  terminates  in  a  vertically  mounted  key  unit  which  has  three  keys 
with  levers  in  a  vertical  row.  Each  key  lever  can  be  operated  either  up  or  down  or  remain 
in  its  normal  middle  position.  The  upper  and  middle  horizontal  rows  of  keys,  when  op- 
erated up  or  down,  connect  their  respective  trunks  or  extensions  to  a  common  strapping 
between  the  keys  in  the  board,  and  the  lower  horizontal  row  of  keys  does  likewise  when 
operated  to  the  up  position.  In  the  down  position,  these  latter  keys  bridge  a  holding  coil 
across  the  trunk  for  the  trunk  keys  and  apply  ringing  current  to  the  extensions  for  the  ex- 
tension keys,  and  the  ringing  keys  are  non-locking. 

Each  trunk  has  a  visual  drop  and  condenser  bridged  across  it,  and  each  extension  has  a 
visual  signal  in  series,  so  that  the  central  office  and  the  extensions  can  each  signal  the  at- 
tendant. A  supervisory  relay  is  provided  in  each  connecting  circuit,  which  circuit  also 
provides  talking  battery  for  extension  to  extension  connections,  through  a  retard  coil. 
For  trunk  to  extension  connections,  talking  battery  is  furnished  over  the  trunk  from  the 
central  office.  The  attendant's  telephone  set  is  connected  to  the  first  vertical  key  unit  on 
the  right  side  of  the  board. 

Nominal  power  of  24  volts  direct  current  is  required  for  talking  battery  between  ex- 
tensions or  between  the  attendant  and  the  extensions  and  is  usually  furnished  over  cable 
pairs  from  the  office,  as  is  the  required  20-cycle  ringing  current  for  the  board.  A  hand 
generator  is  provided  for  the  board  for  emergency  use  or  where  office  ringing  power  is  not 
available. 

Incoming  rings  over  a  trunk  operate  the  trunk  drop,  and  the  attendant  answers  by 
operating  an  idle  connecting  circuit  key  associated  with  the  trunk,  and  also  the  attendant's 


PRIVATE  BRANCH  EXCHANGE  EQUIPMENT        17-115 

corresponding  key,  both  in  the  same  horizontal  row  and  to  the  same  position,  up  or  down. 
This  connects  the  attendant's  telephone  to  the  trunk  line. 

If  an  extension  is  being  called  over  a  trunk,  both  trunk  and  extension  keys  are  operated 
to  corresponding  positions  and  the  extension  ringing  key  is  operated.  When  or  before  the 
extension  answers,  the  attendant  restores  the  attendant's  telephone  set  key  to  normal,  and 
when  the  supervisory  signal  indicates  that  the  conversation  is  finished  both  the  trunk  and 
extension  keys  are  operated  to  normal. 

Extension  to  extension  calls  are  established  similarly  by  properly  operating  the  calling 
and  called  extension  keys. 

Trunk  calls  can  be  held  by  the  attendant  operating  the  trunk  holding  key.  The  at- 
tendant or  any  extension  may  dial  through  the  board  to  a  dial  office,  if  the  telephone  set  is 
equipped  with  a  dial.  Outgoing  trunk  calls  signal  in  to  a  manual  common-battery  office 
automatically. 

Another  type  of  manual  cord  common  battery  PBX  board  has  a  capacity  of  320  extensions, 
15  central  office  trunks,  and  15  cord  circuits.  This  board  is  of  the  two-panel,  single- 
position,  non-multiple  type  suitable  for  medium-size  installations  ranging  from  80  to  320 
extensions.  Two  boards  may  be  operated  side  by  side  to  increase  the  capacity.  The 
trunks,  central  office  or  tie  (to  another  PBX),  and  the  extensions  terminate  on  jacks  be- 
tween which  the  connections  are  established  by  means  of  the  cord  circuits.  The  trunks 
are  ringdown  incoming  to  and  automatic  signaling  outgoing  from  the  PBX.  Each  pair 
of  cords  has  a  talking  and  listening  key  and  a  ringing  key  for  ringing  on  either  the  front  or 
back  cord,  and  double  lamp  supervision,  except  that  supervision  is  provided  on  the  ex- 
tension cord  only,  for  trunk  calls.  Each  trunk  and  extension  circuit  is  equipped  with 
lamp  signals  for  signaling  the  attendant,  and  up  to  20  line  relays  may  be  provided  for  ex- 
tensions to  increase  their  signaling  limit. 

The  attendant's  telephone  circuit  contains  a  dial  for  dialing  on  dial  trunks  or  extensions. 
The  board  may  be  operated  in  conjunction  with  dial  PBX  boards  for  intercepting  calls 
or  other  services. 

Talking  battery  is  supplied  for  extension  to  extension  connections  through  the  single 
bridged  retard  coil  in  each  cord  circuit,  but  for  trunk  to  extension  connections  talking 
battery  is  furnished  from  the  central  office.  Nominal  24-volt  battery  required  at  the  PBX 
for  talking  and  circuit  operations,  and  20-cycle  ringing  power  for  ringing  extensions,  are 
usually  provided  over  cable  pairs  from  the  central  office.  A  hand  generator  is  usually 
furnished  for  emergency  use. 

This  board  is  designed  to  establish  connections  between  local  extensions  and  between 
these  extensions  and  a  manual  or  dial  central  office  or  other  PBX. 

Incoming  calls  from  either  a  trunk  or  extension  to  an  extension  light  an  associated  lamp 
signal  at  the  board,  and  the  attendant  answers  by  operating  the  talking  and  listening  key 
of  the  answering  cord,  which  has  been  inserted  in  the  calling  jack.  The  connection  is  then 
completed  by  inserting  the  calling  cord  in  the  called  extension  jack  and  ringing  this  ex- 
tension. Outgoing  calls  to  the  central  office  are  handled  over  a  trunk  similarly,  except 
that  signaling  the  central  office  is  usually  automatic. 

A  type  of  large  manual  cord  common-battery  multiple  PBX  board  has  a  capacity  of  1520 
extensions  (without  designation  strips),  240  trunks,  and  15  cord  circuits  (per  position). 
The  number  of  extensions  may  be  increased  by  using  a  34  l/2-in.  in  place  of  the  usual 
24  Vs  in.  jack  panel  opening.  There  are  two  panels  per  position  and  four  panels  per 
multiple  jack  appearance  for  both  extensions  and  trunks  and  one  position  per  switchboard 
section. 

The  trunks  and  extensions  terminate  on  series  cutoff  type  jacks  in  the  face  of  the  board, 
and  each  jack  has  a  multiple  line  lamp  (for  manual  operation)  associated  with  it.  Since 
each  extension  or  trunk  appears  in  the  jack  multiple  at  every  fourth  panel  throughout  the 
board,  the  multiple  lamps  can  be  arranged  to  light,  for  an  incoming  call,  at  each  appear- 
ance up  to  a  total  of  four,  thus  attracting  the  attention  of  a  greater  number  of  attendants 
and  improving  answering  performance.  Line  relays  may  be  provided  for  the  extensions 
to  increase  their  signaling  range.  The  manual  trunks  are  usually  ringdown  incoming  and 
automatic  outgoing,  and  for  a  dial  office  they  are  of  the  dialing  type. 

The  cord  circuits  are  of  the  bridged-impedance  series-condenser  type,  which  provide 
battery  feed  to  each  cord  of  the  pair  separately  through  bridged  retard  coils.  The  tip  and 
ring  of  the  cord  pair  each  has  a  condenser  in  series  which  prevents  d-c  flow  hi  one  half  of 
the  cord  circuit  affecting  d-c  flow  in  the  other  half.  Talking  battery  is  thus  supplied  each 
extension  from  the  PBX  cord  circuit  for  extension  to  extension  calls  and  to  the  extension 
for  tie  trunk  to  extension  calls.  For  central-office  trunk  to  extension  calls  the  cord  cir- 
cuit is  so  arranged  that  extension  talking  battery  is  furnished  from  the  central  office,  except 
where  PBX  long  line  circuits  are  provided  at  the  PBX. 

Each  cord  pair  has  a  talking  and  dial  key  and  a  ringing  key  for  ringing  on  either  the 


17-116  TELEPHONY 

front  or  back  cord.  Double  lamp  supervision  is  provided  for  each  cord  circuit  on  extension 
to  extension  calls,  and  single  supervision  for  trunk  to  extension  calls.  A  position  dial  may 
be  cut  in  on  any  cord  for  dialing  by  operating  the  talk  and  dial  key. 

The  distributing  frames  are  enclosed  in  sections  at  the  head  of  the  switchboard.  Fac- 
simile sections  may  also  be  located  in  the  switchboard  line-up.  The  board  may  be  oper- 
ated in  conjunction  with  dial  PBX  boards  for  intercepting  or  other  services. 

The  power  required  for  operation  of  this  board  is  a  nominal  48-volt  d-c  source  for  talk- 
ing and  circuit  operations,  and  20-cycle  ringing  current.  Owing  to  the  usually  relatively 
heavy  battery  drains,  storage  batteries  are  provided  at  the  PBX  and  charged  by  a  separate 
power  unit  at  the  PBX  or,  where  economical  and  practicable,  over  cable  pairs  from  the 
central  office.  Ringing  current  is  also  generally  supplied  to  the  PBX  over  cable  pairs 
from  the  office,  and  each  position  is  equipped  with  a  hand  generator  for  emergency  use. 

This  board  is  designed  for  establishments,  such  as  large  department  stores,  institutions, 
and  organizations,  where  PBXs  of  the  smaller  types  are  not  adequate.  It  may  be  oper- 
ated in  manual  or  dial  office  areas  and  in  conjunction  with  other  manual  or  dial  PBXs. 

Operation  is  similar  to  a  regular  exchange  manual  switchboard  except  for  the  several 
special  features  that  may  be  provided  to  meet  the  individual  subscriber's  needs. 

A  typical  medium-size  dial  PBX  unit  of  the  step-by-step  (SXS)  type  has  a  capacity 
of  up  to  79  extensions,  10  central  office  and  10  attendant's  trunks,  for  two-digit  dialing. 
The  SXS  equipment  consists  principally  of  20  line  finders,  100  line  and  cutoff  relays,  and 
miscellaneous  equipment  mounted  on  the  side  of  the  switch  frame,  and  20  selector-con- 
nectors and  19  trunk  equipments  mounted  on  the  reverse  of  the  frame. 

This  PBX  unit  functions  with  a  suitable  companion  manual  PBX  containing  a  jack 
appearance  for  each  trunk  and  extension  for  the  purpose  of  receiving,  originating,  and 
intercepting  calls  that  cannot  be  handled  by  the  SXS  unit.  Incoming  calls  reach  the 
manual  attendant,  who  dials  the  called  extension  over  an  attendant's  trunk.  Outgoing 
calls  may  be  handled  and  calls  requiring  special  treatment  are  usually  handled  at  the  man- 
ual board.  The  SXS  unit  handles  direct  calls  between  extensions  and  from  extensions  to 
the  central  office,  associated  manual  PBX  board,  and  other  PBX  boards  equipped  to  re- 
ceive dialing  pulses. 

Power  requirements  for  the  SXS  unit  consist  of  nominal  48  volts  direct  current  for 
talking  and  circuit  operations,  and  machine  ringing  current.  D-c  power  is  usually  sup- 
plied by  a  small  power  unit  located  at  the  PBX,  or  over  central-office  cable  pairs  if  the 
load  permits  and  pairs  are  available.  Ringing  current  is  generally  furnished  over  central- 
office  cable  pairs. 

Dialed  calls  are  completed  by  the  usual  step-by-step  processes  (described  elsewhere  in 
this  section) .  Calls  handled  at  the  manual  PBX  may  either  be  dialed  in  dial  office  areas 
or  completed  over  automatic  signaling  trunks  in  manual  office  areas. 

Dial  PBX  units  of  the  step-by-step  type  having  capacities  up  to  3200  extensions,  depend- 
ing upon  the  number  of  extensions  required,  are  available.  Two-digit  dialing  (under  100 
extensions)  requires  line  finders  and  selector-connectors  (operating  as  a  selector  on  the 
upper  levels  and  a  connector  on  the  lower  levels)  while  four-digit  dialing  requires  first  and 
second  selectors  and  connectors.  If  incoming  dial  repeating  trunks  are  provided,  they 
terminate  on  incoming  connectors  for  two-digit  and  on  incoming  selectors  for  three-  or 
four-digit  dialing.  The  line  finders  have  a  200-point  bank;  the  other  switches  have  100- 
point  banks. 

These  dial  PBX  units  are  usually  associated  with  manual  PBX  units,  the  latter  receiving, 
originating,  and  intercepting  calls  that  the  SXS  unit  is  not  arranged  to  handle.  The  SXS 
unit  handles  calls  between  extensions  and  calls  to  the  central  office  or  to  another  PBX 
which  are  capable  of  receiving  dial  pulses  from  the  extensions  or  over  dial  repeating  trunks. 
The  extensions  of  the  SXS  unit  may  be  reached  direct  from  other  PBXs  if  proper  dialing 
facilities  are  provided,  but  calls  to  these  extensions  from  the  central  office  are  usually 
handled  at  the  manual  PBX  board. 

PROTECTION  for  PBX  trunks  and  extensions  at  the  PBX  is  the  same  as  for  regular 
subscriber  to  central-office  lines  of  the  same  classification,  that  is,  exposed  or  non-exposed. 
In  addition,  owing  to  paths  to  ground  through  the  equipment  in  the  PBX  board,  which 
may  permit  foreign  currents  large  enough  to  cause  trouble  to  flow  through  the  equipment, 
heat  coil  type  fuses  having  the  same  operating  characteristics  as  central-office  heat  coils 
(except  that  they  open  rather  than  ground  the  line  wires)  are  provided  in  each  exposed 
trunk  and  extension  at  the  PBX.  In  the  case  of  large  PBXs  a  section  of  fine-gage  cable 
sometimes  is  used  in  place  of  the  7-ampere  line  fuses.  This  is  similar  to  the  fusing  pro- 
vided at  central  offices  (see  article  6). 

ORDER  TURRETS,  manufactured  in  various  types  and  capacities,  are  employed  in 
business  or  service  establishments,  such  as  department  stores,  telegraph  offices,  taxicab 
companies,  and  newspaper  offices,  to  receive  orders  for  goods  or  requests  for  services. 


PRIVATE  BRANCH  EXCHANGE  EQUIPMENT        17-117 

One  type  of  order  turret  for  multidepartment  businesses  has  a  capacity  of  one  two-way 
trunk  and  one  outgoing  trunk  from  the  turret  to  the  subscriber's  PBX  and  one  incoming 
trunk  from  the  PBX  to  the  turret.  The  incoming  customer  calls  are  received  at  the  PBX 
board  and  routed  over  trunks  to  the  proper  department  turret,  where  an  attendant  re- 
ceives the  customer's  request.  The  turret  serves  as  a  small  switching  unit  between  a 
single  telephone  in  the  departments  thus  equipped  and  the  PBX,  and  it  avoids  the  in- 
stallation of  several  telephones  at  each  location. 

An  order  turret  system,  where  incoming  calls  are  concentrated  in  one  line-up  of  turret 
positions,  is  also  available.  Incoming  customer  calls  from  either  a  manual  or  dial  central 
office  or  PBX  are  received  in  incoming  trunk  circuits,  allotted  by  an  allotter  circuit  to  a 
sequence  storing  circuit,  and  thence  released  to  an  idle  attendant  in  the  sequence  in  which 
the  calls  are  received.  Thus  the  calls  are  distributed  automatically  as  rapidly  as  turret 
attendants  become  idle  and  with  mimmum  loss  of  time  in  handling.  This  type  of  system 
has  a  capacity  of  120  incoming  trunks  (with  first  group  of  selectors),  110  attendant's  trunk 
circuits,  and  110  attendant's  positions.  A  small  order  turret  located  in  front  of  each  at- 
tendant has  four  keys  and  two  lamps,  a  trunk,  and  a  calling  waiting  signal  for  receiving 
and  originating  calls.  Calls  can  be  held,  released,  or  transferred  bjr  means  of  these  keys. 
Usually  a  manual  or  combined  manual  and  dial  PBX  board  is  associated  with  this  turret 
system.  Power  supply  is  20-28  volts  direct  current,  for  which  a  local  automatic  power 
plant  is  provided.  Hinging  current  is  usually  furnished  over  cable  pairs  from  the  central 
office. 


SECTION  18 
TELEGRAPHY 


THEORY 
ART.  -^T  J°HN  D.  TAYLOR  PAGB 

1.  Methods  of  Transmission 02 

2.  Codes 02 

3.  Telegraph  Signals 05 

4.  Wave  Shapes 05 

5.  Distortion 11 

TELEGRAPH  SYSTEMS 
BY  JOHN  D.  TAYLOR 

6.  Direct-current  Systems 18 

7.  Automatic  Telegraph  Systems 26 

8.  Alternating-current  Telegraph  Systems.  35 

9.  Fac3imile  System 38 

10.  Miscellaneous  Transmitting  and  Signal- 

ing Systems 38 

SUBMARINE  CABLE  TELEGRAPHY 
BY  JOHN  D.  TAYLOR 

11.  Cable  Data 41 

12.  Operation 42 


TELEGRAPH  EQUIPMENT 
Axrm  BY  JOHN  D.  TAYLOB  PAGm 

13.  Central-office  Equipment 46 

14.  Station  Equipment 51 

TRANSMISSION-MAINTENANCE 
BY  JOHN  D.  TAYLOR 

15.  Transmission  Standards 53 

16.  Transmission  Coefficients 54 

17.  Crossfire 54 

18.  Maintenance 55 

RADIO  TELEGRAPH  SYSTEMS 
BY  J.  L.  FINCH 

19.  Choice  of  Transmitter  and  Receiver  Sites  56 

20.  Choice  of  Frequencies 56 

21.  Reduction  of  Fading  Effects 57 

22.  Radio  Interference 58 

23.  Frequency  Shift  Keying 58 

24.  Traffic  Office  and  Equipment 5S 

25.  Control  Channels 60 


18-01 


TELEGRAPHY 
THEORY 

By  John  D.  Taylor 

Fundamentally,  the  process  of  communicating  by  telegraph  consists  of  sending  electrical 
impulses  by  wire  or  radio  from  a  transmitting  to  a  receiving  point.  These  impulses  are  so 
selected,  arranged,  and  transmitted  (in  sequence) ,  that  they  are  received,  interpreted,  and 
recorded  as  intelligible  characters  at  the  receiving  point.  Such  interpretation  may  be 
made  by  the  ear  listening  to  the  click  of  a  telegraph  sounder  or  the  tone  in  a  head  receiver 
or  loudspeaker,  or  by  automatic  and  mechanical  devices. 

Basically,  it  was  necessary  to  develop  codes  (intelligence-bearing  arrangements  of  elec- 
trical impulses)  and  to  provide  transmitting  and  receiving  devices  and  interconnecting 
channels,  capable  of  satisfactorily  passing  these  codes  between  the  desired  locations. 

1.  METHODS  OF  TRANSMISSION 

Telegraph  codes  or  signals  are  transmitted  electrically  (1)  by  open-wire  land  lines  or 
cables  or  short  submarine  cables,  using  either  direct  or  alternating  current,  (2)  by  long 
submarine  cables  (oceanic),  using  only  direct  current,  and  (3)  by  radio,  using  only  alter- 
nating current. 

Direct-current  telegraphy  employs  a  d-c  source  of  power,  which  is  either  interrupted, 
reversed  in  polarity,  or  altered  in  magnitude  by  the  transmitting  mechanism.  The  line 
current  frequency  with  present  high-speed  telegraph  systems  does  not  usually  exceed  about 
200  cycles  per  second,  and  thus  systems  using  this  type  of  transmission  are  generally 
referred  to  as  low-frequency  systems. 

Alternating-current  telegraphy  employs,  for  each  transmitting  channel,  a  relatively 
narrow  band  (about  170  to  300  cycles  wide)  of  a-c  frequencies,  which  are  modulated  by 
the  telegraph  signals,  being  transmitted,  before  being  applied  to  the  line.  A  basic  block 
of  frequencies,  approximately  3000  cycles  wide,  is  used  to  provide  a  group  of  carrier  chan- 
nels, the  number  of  channels  per  block  depending  on  the  width  allocated  to  the  individual 
channels.  Carrier  telegraph  systems  utilize  one,  two,  four,  or  more  basic  blocks  of  fre- 
quencies in  each  direction  of  transmission,  the  blocks  being  stacked  one  above  the  other  in 
the  assigned  frequency  range  by  translation  or  second-modulation,  as  discussed  in  article  3, 
Section  17. 

The  transmission  and  reception  of  signals  may  be  either  manual  or  automatic.  In 
manual  operation,  the  transmitted  signals  are  formed  by  a  hand-operated  switch  or  sending 
key,  and  the  signals  are  received  either  on  an  electromagnetically  operated  sounder,  which 
converts  the  signals  into  audible  clicks,  or  on  a  Morse  recorder,  which  makes  an  ink  record 
of  the  received  impulses.  These  audible  or  recorded  signals  are  interpreted  by  an  opera- 
tor, who  typewrites  the  message  manually.  Automatic  operation  provides  for  the  forma- 
tion and  transmission  of  the  signals  by  an  automatic  circuit-interrupting  device  or  trans- 
mitter, which  may  be  controlled  either  directly  or  indirectly  by  a  group  of  keys  similar  to 
a  typewriter  keyboard.  The  received  signals  are  automatically  interpreted  and  recorded 
in  typewritten  form  by  a  printer.  In  some  cases,  notably  in  the  operation  of  submarine 
cables,  a  combination  of  automatic  transmission  and  manual  reception  is  employed. 
The  transmitting  and  receiving  equipment,  and  the  associated  line  terminal  apparatus, 
are  so  closely  related  that  the  entire  assembly  is  usually  referred  to  as  a  telegraph  system. 

2.  CODES 

Characters  of  the  alphabet,  figures,  and  punctuation  marks  are  transmitted  as  combina- 
tions of  marking  and  spacing  impulses.  A  marking  signal  or  impulse  is  one  which  causes 
the  receiving  apparatus  to  be  operated,  and  a  spacing  signal,  which  separates  successive 
marking  signals,  places  the  receiving  apparatus  in  the  unoperated  condition.  Spacing 
signals  may  be  intervals  of  no  current,  impulses  of  opposite  polarity  to  the  marking  sig- 

18-02 


CODES  18-03 

nals,  or  impulses  of  different  current  value  from  the  marking  signals.  Two-element  codes 
employing  only  two  conditions  of  current,  positive  and  negative,  or  current  and  no  cur- 
rent, are  mainly  used  on  land  line  circuits.  Three-element  codes,  which  use  both  positive 

Telegraph  Codes 
Morse  Continental  Cable  Morse 

...............      •-  •-  4--o 


-++4-0 
-4—  4-0 

...............  -'  '  --  •  -+4-0 

E  ...............  •  •  4-0 

£  ...............  •-•  -•-•  ++-4-0 

g  ...............  ---  ---  --4-o 

?  ...............  -  -  -  -  -  -  -  -M--H-0 

\  ...............  '  '  •   •  4-+0 

J  ...............  ----  ----  4-  ---  0 

K  ...............  ---  ---  -+-  o 

*V  ..............  —  -  -•  •  4—  4~ho 

M  ..............  --  --  --  0 

N  ...............  -  .  _.  _  +  0 

O  ...............  -      .  ---  ---  0 

P  ...............  .....  -  ---  4—  -4-0 

Q  ...............  .._.  ---  -  ---  1  —  o 

R  ...............  •    •  -  -  --  4—  4-o 

S  ...............  -  •  -  -  -  -  4-4-4-0 

T  ...............  -  -  -0 

U  ...............  -  -  -  •  -  -  4-4—0 

v  ...............  .  .  .  _  .  .  .  _  4-4-4—  o 

W  ..............  •  --  -  --  4  ---  o 

X  ...............  ---   -  -.._  _4-_{—  Q 

Y  ...............  .  .    .:  -  .  -_  —  1---0 

Z  ...............  ....  ---    .  _ 


1  ...............        ----  -----  4-  ----  o 

2  ...............  .._..  ..  ---  ++  ---  0 

3  ...............  ..._.  .    .    .  __ 

4  ...............  .    .    .    .  _  ...... 

5  ...............  ---  .....  4-4-4-+4-0 

6  ...............  ......  _  .    .    .    .  ^^^+4.0 

7  ...............        ---  -  ---  -  -  —  4-4-4-0 

8  ...............         -....  ----    .  ---  +4-0 

9  ...............          _..-  -----  ----  4-0 

o  ...............       -  -----  -----  o 

Period  ...........  -    ----    •  ......  ++     ++     ++ 

Comma  ..........  .  —  .  —  ._._._  -}        |       j 

Semicolon  ........  .....  _._._.  —  j        j        f- 

Colon  ...........  ---      •       •  ---  *    '    '  ----  l"j-  + 

Interrogation  .....  —...—.  •    •  --  -    -  4    ^7"~"rj~ 

Quotation  ........  .._.       —  .  ._.._.  •]        r-j        j- 

Short  figures  used  in  Continental  and  Cable  Morse  Codes  where  no  confusion  would  result'. 

1....  -  -                                     6.... 

2....  •   -  -                                 7....          ---   -   • 

3..,.  -   -    ---                         8..-- 

4=....  ....  -                         9.... 

5....  -                                          0.... 

FIG.  1.  Telegraph  Codes—  Morse,  Continental,  and  Cai>le  Motse 

and  negative  polarities  and  also  a  zero  or  no-current  interval  to  separate  groups  of  im- 
pulses, are  employed  chiefly  on  long  non-loaded  submarine  cables,  where  the  comparative 
freedom  from  extraneous  interference  removes  the  chief  objection  to  the  use  of  a  zero 
interval  for  the  separation  of  signal  groups. 

The  Morse  Code  is  used  Almost,  exclusively  in  the  United  States  on  hand-worked  land 
lines;  the  Continental  Code  has  been  adopted  by  almost  all  foreign  telegraph  administra- 


18-04 


TELEGRAPHY 


submarine  cables. 


Characters 


Code  signals- 


tions,  and  is  universally  employed  on  radio  telegraph  circuits.  In  these  two  codes,  the  dot 
signal  is  the  basis  of  time  measurement.  A  dash  is  three  times  the  length  of  a  dot;  the 
spaces  separating  successive  impulses  in  a  combination  are  of  dot  length;  the  spaces  be- 
tween impulse  groups  are  equal  to  three  dots;  and  a  space  equivalent  to  six  dots  is  used 
to  separate  words.  The  Cable  Morse  Code  is  used  almost  entirely  in  the  operation  of  long 
All  impulses  and  spaces  are  of  equal  length,  a  positive  impulse  repre- 
senting a  dot  and  a  negative  impulse  repre- 
senting a  dash. 

In  calculating  speeds,  the  average  Morse 
character  is  equivalent  in  length  to  approxi- 
mately 8.5  impulses  of  dot  length,  or  about 
4.25  cycles,  while  Continental  characters 
average  about  9  impulses  or  4.5  cycles  each. 
The  Cable  Code  averages  3.7  impulses  or 
1.85  cycles  per  letter.  These  figures  take 
into  account  the  frequency  with  which  the 
various  letters  of  the  alphabet  occur  in 
ordinary  telegraph  traffic. 

Figure  1  shows  the  combinations  of  dot 
and  dashes  used  in  the  Morse  and  Conti- 
nental Codes,  and  the  positive  and  negative 
polarity  and  zero  current  intervals  used  in 
the  Cable  Morse  Code. 

While  the  Morse  and  Continental  Codes 
are  designed  for  manual  operation,  the 
Cable  Morse  Code  is  transmitted  automati- 
cally. The  automatic  transmission  of  tele- 
graph signals  is  almost  universally  employed 
for  land  line,  cable,  and  radio  telegraph  sys- 
tems, because  of  its  greater  speed  and  ease 
of  operation,  resulting  in  large  economies. 
Manual  operation,  when  and  where  used,  is, 
in  general,  confined  to  short-haul  traffic  and 
special  services  of  such  a  nature  as  not  to 
warrant  or  be  adaptable  to  automatic 
operation. 

In  developing  automatic  telegraph  sys- 
tems, it  was  necessary  to  design  machines  or 
devices,  which  would  automatically  send 
and  receive  electrical  impulses  (positive  or 
negative)  and  to  devise  individual  arrange- 
ments of  these  impulses,  representing  intelli- 
gible characters.  It  was  further  required 
that  these  arrangements  could  be  set  up 
manually  for  transmission,  and  sent,  re- 
ceived, and  interpreted  automatically  at 
speeds  suitable  to  the  transmitting  medium. 
One  type  of  sending  and  receiving  device 
for  use  in  automatic  systems  is  the  teletype- 
writer or  teleprinter  (described  in  detail  in  article  7  of  this  section),  which  utilizes  a  5-im- 
pulse  code  for  each  character  with  a  single  impulse  for  starting  and  also  for  stopping  both 
sending  and  receiving  units  between  characters.  The  code  devised  for  the  teletypewriter, 
which  is  practically  identical  with  that  of  the  teleprinter,  is  shown  in  Fig.  2,  in  which  L.C. 
and  U.C.  are  abbreviations  for  lower-  and  upper-case  characters  on  the  typing  keyboard 
and  the  black  and  white  spaces  are  the  marking  and  spacing  impulses,  respectively.  The 
signal  impulse  lengths  in  milliseconds  (ms)  are  also  shown  for  different  operating  speeds 
(words  transmitted  per  minute) . 

Of  the  32  separate  combinations  available,  using  5  equal-length  impulses  for  each 
character,  the  combination  of  all  spacing  impulses  has  no  character  assigned  to  it  and  is 
not  transmitted. 

The  line  frequency  for  the  teletypewriter  operation  depends  upon  the  number  of  words 
per  minute  being  transmitted.  For  60-speed  transmission  (about  60  words  per  minute) 
the  shortest  signal  element  is  about  0.022  sec  (see  Fig.  2) ,  and  the  line  frequency  is 

I/ (0.022 X  2)  =  22.7  cycles  per  second 


Signal  lengths  In  milliseconds  Standard  speed 


FIG. 


Teletypewriter   Code 

System) 


(Courtesy   Bell 


WAVE  SHAPES 


18-05 


3.  TELEGRAPH  SIGNALS 

Electrical  impulses  must  be  formed  by  the  transmitting  device,  transmitted  through  a 
connecting  medium,  and  finally  received  and  interpreted  by  the  receiving  device,  so  that 
the  received  message  is  identical  with  the  original  sent  message  and  is  efficiently  trans- 
mitted. However,  owing  to  the  electrical  characteristics  of  telegraph  circuits  and  asso- 
ciated Apparatus,  telegraph  signal  currents  are  generally  more  or  less  modified  (electri- 
cally) in  transmission,  and,  if  suitable  corrections  were  not  made,  these  modifications  or 
distortion  would,  in  many  cases,  cause  errors  in  the  received  message. 

Telegraph  signals,  in  d-c  operation,  are  classed  as  (1)  neutral  (current  flows  over  the 
line  in  either  direction  for  the  operated  or  marking  position  and  no  current  flows  for  the 
non-operated  or  spacing  position  of  the  line  relays)  or  (2)  polar  (current  flows  over  the 
line  in  one  direction  for  the  marking  and  in  the  opposite  direction  for  the  spacing  position 
of  the  line  relays) .  For  either  type  of  signal,  the  change  of  current  from  mark  to  space  or 
space  to  mark  is  known  as  a  transition. 


Time 
A 


4.  WAVE  SHAPES 

NEUTRAL  SYSTEM.  The  change  in  line  current  values,  with  respect  to  time,  may 
be  plotted  to  show  the  wave  shape  of  the  telegraph  signal  for  any  telegraph  circuit,  as  shown 
in  Fig.  3  for  a  neutral  circuit  transmitting  the  Morse  signal  (dot,  dash)  representing  the 
letter  A.  For  such  a  circuit,  A  in 
Fig.  3  shows  the  wave  shape  of  the 
current,  if  the  times  of  building  up 
and  decaying  to  the  steady-state 
values  are  neglected,  B  shows  the 
effect  on  the  wave  shape  of  series 
inductance  (line  relay  or  composite 
set  winding) ,  and  C  shows,  by  the 
shaded  area,  the  additional  effect 
on  the  wave  shape  of  the  composite 
set  or  other  condensers. 

With  reference  to  C,  Fig.  3,  when 
the  telegraph  sending  key  closes, 
the  inductance  in  the  circuit  op- 
poses any  sudden  change  in  current 
value,  and  part  of  the  current  is 
diverted  from  the  line  to  charge  the 
composite  set  condensers  to  about 
the  applied  battery  potential,  as 
the  line  current  builds  up  to  its 
maximum  value;  when  the  key 
opens,  the  current  immediately 
starts  to  decay  and,  although 


r\  r 

\ 

Time 
0 

FIG.  3.   Wave  Shape  for  Letter  A — Neutral  Telegraph  System 
(Courtesy  Bell  System) 


partly  sustained  by  flow  of  current  from  the  condensers  and  the  opposition  of  the  induct- 
ance to  current  change,  shortly  reaches  its  minimum  value. 

Current  wave  shapes  are  important  factors,  which  affect  telegraph  relay  performance 
and  adjustments.  For  any  given  relay,  there  is  a  definite  operating  and  release  (less  than 
the  operating)  current  value  for  given  operating  conditions.  Figure  4  shows  a  schematic 
d-c  circuit,  containing  a  neutral  type  of  telegraph  relay,  milliarn meter,  battery,  and  rheo- 
stat. The  retractive  spring  tension  is  controlled  by  adjusting  screw  Si,  the  armature  to 
pole  piece  air  gap  by  S2,  and  the  armature  travel  distance  by  83  and  £4. 

Figure  5  shows  the  effect  of  relay  adjustments  on  operating  and  release  current  values 
(indicated  by  the  black  dots,  0,  R,  0i,  and  RI  on  the  curves),  and  on  the  effective  length 
(T  and  TI)  of  the  telegraph  signal.  In  telegraph  parlance,  the  shorter  marking  signals 
are  called  "light"  and  the  longer  marking  signals  "heavy."  In  practice,  relay  adjust- 
ments are  made,  as  required,  to  provide  satisfactory  received  signals  from  usually  dis- 
torted wave  shapes,  caused  by  the  electrical  or  mechanical  characteristics  of  the  line  and 
associated  equipment  or  changes  therein. 

Effective  signal  length  might  also  be  increased  or  decreased  for  a  given  neutral  tele- 
graph circuit  with  a  fixed  relay  adjustment  by  raising  or  lowering,  respectively,  the  applied 
circuit  voltage.  However,  since  line  current  values  are  limited,  in  practice,  by  crossfire 


18-06 


TELEGRAPHY 


into  other  telegraph  circuits  or  by  interference  with  telephone  transmission,  this  method 
of  controlling  the  signal  length  is  restricted  and  not  usually  employed. 


FIG.  4.     Schematic  of  a  0-c  Telegraph  Circuit  with  Neutral  Relay  (Courtesy  Bell  System) 

Relay  adjustments  are  also  affected  by  line  conditions,  such  as  leakage  of  line  currents 
to  ground  through  tree  or  other  contacts  or  the  direct  capacitance  between  the  line  wire 
and  ground.  In  general,  for  open-wire  telegraph  circuits  this  leakage  factor  varies  as  among 

*  Relay  operating1  points 


TJme 
A 


lime 
B 


FIG.  5.    Effect  of  Relay  Adjustment  on  Telegraph  Relay  Operation  (Courtesy  Bell  System) 

wires  and  increases  materially  from  the  dry  to  wet  weather  condition.  Figure  6a  shows  a 
neutral  telegraph  circuit  with  grounded  battery  at  one  station  and  Fig.  66  the  same  cir- 
cuit with  grounded  battery  at  both  stations,  and  for  both  circuits  the  leakage  to  ground, 


FIG.  6.     Neutral  Telegraph  Circuits  with  Leakage  to  Ground  (Courtesy  Bell  System) 

which  may  be  concentrated  principally  at  one  or  more  points  or  generally  distributed  along 
the  line,  is  represented  by  the  resistance  S. 


WAVE  SHAPES 


18-07 


Figure  7  shows  leakage  effects  on  the  current  curve  for  the  circuit  (Fig.  6a).  The  curve 
in  Fig.  7,  A,  represents  the  ideal  condition  of  no  leakage.  The  curve  in  B  represents  the 
leakage  current  alone  through  S.  Since  the  inductance  through  S  is  less  than  through 
station  B,  the  leakage  curve  in  B  is  steeper  than  the  curve  in  _-L,  and  this  resultant,  which 
affects  station  A  relay,  may  be  about  as  shown  in  C.  In  this  curve,  the  effective  signal 
length  Tc  is  greater  (heavier  signal)  than  TamA.  The  leak  S  also  shunts  the  path  through 
station  B  and  tends  to  decrease  the  line  current  to  this  station,  as  well  as  to  flatten  the 
current  curve,  as  shown  by  D. 

The  wide  variation  of  effective  signal  lengths,  Tc  and  Td,  through  station  A  and  B  relays, 
in  series,  would  not  occur  if  half  of  the  total  line  battery  voltage  was  applied  at  each  sta- 
tion, oppositely  poled  to  the  line,  as  shown  in  Fig.  66.  Since  each  battery  supplies  current 
to  the  leakage  path  £,  but  in  oppo- 
site directions,  the  resultant  current 
through  this  path  will  generally  be 
small  and  the  line  current  through 
the  station  A  and  B  relays  will  be 
more  nearly  equal  and  stabilized 
than  for  single  end  battery  feed,  thus 
improving  signal  transmission. 

Telegraph  current  wave  shapes 
formed  by  direct  current  are,  in 
effect,  somewhat  similar  to  a-c  wave 
shapes  and  contain  various  harmon- 
ics of  the  fundamental  frequency,  as 
will  be  discussed  later.  Owing  to 
the  normal  line  wire  capacitance  to 
ground  (irrespective  of  other  leakage 
paths),  the  a-c  components  of  the 
telegraph  currents  will  be  shunted,  in 


X 


Time 
A 


Time 
B 


Time 
C 


Time 
D 


FIG.  7.    Effects  of  Leakage  on  Current  Curves  for  the  Neu- 
tral Telegraph  Circuit  of  Fig.  6a  (Courtesy  Bell  System) 


some  degree.  Thus,  the  line  capaci- 
tance not  only  reduces  the  effective 
current  at  the  receiving  station  but 
also  tends  to  distort  the  current 
wave  shape  and  to  limit  the  length  of  operating  line  section  between  repeater  points. 

As  indicated  in  Fig.  5,  the  relay  operating  points  occur  on  a  current  curve  between 
points  of  transition.  The  change  from  the  spacing  to  the  marking  condition  is  designated 
space-to-mark  (S-M]  transition,  and  the  change  from  the  marking  to  the  spacing  condition 
is  designated  mark-to-space  (M-S)  transition.  At  the  sending  end  of  a  telegraph  circuit, 
the  S-M  transition  takes  place  when  the  key  is  closed,  and  the  M-S  transition  occurs  when 
the  key  is  opened. 

There  is  a  certain  time  delay  from  the  closing  of  the  sending  key  to  the  operation  of  the 
receiving  relay  and  from  the  opening  of  the  key  to  the  release  of  the  relay.  In  the  first 
case,  this  delay  is  designated  space-Jo-mark  transition  delay  (S-MTD)  and  in  the  second 
case,  this  delay  is  designated  mark-to-space  transition  delay  (M-STD},  as  shown  in  Fig.  86. 

In  Fig.  8a  the  line  capacitance  is  represented  by  the  dotted  condenser  path  to  ground. 
When  the  sending  key  is  closed,  this  capacitance  is  charged  by  current  flowing  over  the 
line  from  the  sending  station,  and  the  current  is  retarded  in  building  up  to  its  full  value 
by  the  inductance  in  the  circuit,  as  indicated  by  the  sloping  wave  shape  in  Fig.  S6.  The 
current  which  flows  continuously  in  the  biasing  winding  of  the  receiving  relay  produces  a 
constant  magnetic  field  tending  to  hold,  in  this  particular  circuit,  the  receiving  relay 
armature  against  its  spacing  contact.  When  a  marking  signal  current  is  received,  the 
stronger  opposing  magnetic  field  set  up  by  the  larger  marking  current  in  the  operating 
winding  of  the  receiving  relay  causes  the  relay  armature  to  move  to  its  marking  contact. 

Although  the  operating  points  of  a  relay  depend  upon  its  design  and  adjustments, 
it  may  be  assumed,  for  the  purposes  of  this  discussion,  that  the  relays  in  Fig.  Sa  will 
operate  on  33  and  release  on  27  ma  of  direct  current,  or  plus  and  minus  3  ma  from  the 
30  ma  of  biasing  current,  as  shown  in  Fig.  86.  When  the  operating  current  increases  to 
30  ma,  the  effective  operating  current  in  the  receiving  relay  is  then  zero,  but  when  it 
reaches  33  ma,  operation  occurs  and  continues  until  the  line  current,  decaying  on  open 
circuit,  decreases  to  27  ma,  when  the  relay  releases. 

The  S-MTD  period  is  the  time  it  takes  for  the  operating  current  to  increase  from  zero 
to  33  ma,  and  the  M-STD  period  is  the  time  it  takes  for  the  current  to  decay  from  its  max- 
imum value  of  60  ma  to  the  release  value  of  27  ma.  These  periods  vary  from  a  fraction 
of  a  millisecond  (ms)  to  several  milliseconds,  being  determined,  for  a  given  circuit  and 
adjustments,  solely  by  the  circuit  characteristics.  For  a  given  circuit  and  operating  condi- 


18-08 


TELEGRAPHY 


tions,  each  transition  delay  (S-MTD)  will  be  the  same  for  repeated  signals,  and  the  same 
holds  true  for  each  delay  (M-STD) ,  but  the  S-MTD  delays  may  not  be  equal  to  the  M-STD 
delays. 

Each  mark,  of  whatever  length,  begins  with  an  S-M  transition  and  ends  with  an  M-S 
transition.     The  S-MTD  period  reduces  and  the  M-STD  period  increases  the  effective 


Current  In  mils 

o  S  g  £ 

1TD-> 

*~      ~H 

^-                     -»• 

«-S-MTD                            -J 

<-M-STD 

/ 

\ 

/ 

\ 

Relay  bias 

f 

\ 

*v 

f 

V  |  currents* 

TV 

40  60  80  100  120  140 

Time  in  milliseconds 


FIG.  8.     (a)  Neutral  Telegraph  Circuit  and  (&)  Signal  Wave  Shape  (Courtesy  Bell  System) 

length  of  the  mark,  so  that,  if  these  two  delay  periods  are  equal,  the  length  of  the  mark 
will  not  be  changed  by  transmission  over  the  circuit.  Each  space,  of  whatever  length, 
begins  with  an  M-S  transition  and  ends  with  an  S-M  transition.  The  M-STD  period 
reduces  and  the  S-MTD  period  increases  the  length  of  the  space,  so  that,  if  these  two 
delay  periods  are  equal,  the  length  of  the  space  will  not  be  changed  by  transmission  over 
the  circuit.  The  transmission  of  signals  is  considered  perfect  if  the  received  effective 
marks  and  spaces  are  exactly  the  same  length  as  the  sent  marks  and  spaces. 

POLAR  OPERATION.    Wave  shapes  in  polar  telegraph  systems  are  affected  by  circuit 
inductance,  capacitance,  and  leakage  somewhat  as  in  neutral  telegraph  systems.    Figure  9 


Central  office 


Outlying  point 


Line 

\AA/v — i — V\A/v — 


T 


HI; 


•±  -130V 
FIG.   9.      Simplified   One-way  Polar  Telegraph   Circuit    (Courtesy  Bell  System) 

shows  a  simple  one-way  polar  telegraph  circuit,  arranged  to  send  — 130  and  + 130  volt 
impulses  from  the  central  office  (sending  point)  to  an  outlying  receiving  point,  having 
series  line  resistance  and  capacitance  between  the  line  and  ground. 

Assuming  that  the  sending-end  connections  are  adjusted  to  provide  normally  steady- 
state  line  currents  of  +35  ma  (marking)  and  —35  ma  (spacing),  as  shown  in  Fig.  10,  the 
line  capacitance  to  ground  will  delay  the  change  of  line  current  from  spacing  to  marking 
(S-M  transition)  and  from  marking  to  spacing  (M-S  transition) . 


WAVE  SHAPES 


18-09 


The  M-S  and  S-M  wave  shapes  (Fig.  10)  are  identical  in  form  and  symmetrically  located 
about  the  zero  line,  and  the  S-MTD  and  M-STD  are  equal.  Thus,  since  the  sending  end 
potentials  are  equal  and  of  opposite 
sign,  since  the  circuit  resistance  re- 
mains constant  for  both  positions 
of  the  sending  relay  armature,  and 
since  the  operating  points  of  the  g  20 
receiving  relay  are  symmetrically  g. 
located  at  about  the  middle  of  the  1  10 
wave  shapes,  the  received  polar  i  0 
signals  are  unbiased.  | 

TELETYPEWRITER     AND    ^  -10 
TELEPRINTER  OPERATION.   In    |  __ 
teletypewriter  or  teleprinter  opera-    <§    20 
tion,  there  are  five  equal  length  im-       _3o 
pulses,  each  of  which  may  be  nega- 
tive (mark)  or  positive  (space),  in        **40 
accordance  with  the  code  of  signals 
(Fig.  2)  for  this  type  of  operation. 


2O  30  4O 

Time  in  milliseconds 


Telegraph 

character  to  be  transmitted.    Two 

additional  impulses  are  sent  with  each  character,  one  starting  and  one  stopping  the 

machines. 

If  a  series  of  impulses,  representing,  for  example,  the  letter  £>,  are  plotted  with  time  as 
the  horizontal  axis,  as  shown  in  Fig.  11,  it  will  be  noted  that  the  wave  shape,  in  the  upper 


Start     12345       Stop 
Teletypewriter  D  Signal  at  368 

operations  per  minute 


a  D-cr  Component 


b  First  Harmonic  6.1 


c  Second  12.3. 


J 


A    A 


d  Third  18.4 


\ 


e  Fourth  24.5 

r\     r\    r\ 


/Fifth  30.7  - 


\_7 


g  Sixth  36.8 


h  Seventh  43.0 


i  Eignth  49.1 


J  Ninth  55.2 

A    /\    s\  y\    /*\ 


I  D-c  Component-t-firet  harmonic 
(a-i-fc)0-6.1 


Ci-f c)  0-12,3 


V 

n  (wi-HZ)  0-18.4 


o  (w-4-e)  0-24.5 


p  (0-t-/)  0-30.7 


k  Tenth  61.4 
FIG.  11.    Analysis  of  Wave-shape  Components  of  Teletypewriter  Character  D  (Courtesy  Bell  System) 


18-10 


TELEGRAPHY 


left-hand  corner  of  the  figure,  which  is  impressed  on  the  line  at  the  transmitting  end  of 
the  circuit,  has  "square"  corners.  However,  if  such  a  wave  is  analyzed,  it  will  be  found  to 
contain  an  infinite  number  of  component  frequencies,  all  of  which,  in  practice,  cannot  and 
need  not  be  transmitted  to  the  receiving  terminal. 

Though  a  very  simple  receiving  device  might  be  used,  if  all  the  components  could  be 
faithfully  transmitted  over  a  circuit,  such  a  circuit  could  not  be  economically  provided. 
Also,  with  an  ideal  receiving  device,  it  would  only  be  necessary  to  transmit  a  maximum 
frequency  equivalent  to  that  which  considers  the  duration  of  each  unit  signal  element  of 
the  code  as  1/2  cycle  (1  cycle  would  be  the  time  involved  from  the  end  of  element  1  to  the 
end  of  element  3  of  the  square  wave  in  the  upper  left-hand  corner  of  Fig.  11).  This  con- 
dition is  closely  approached  for  long  submarine  cable  telegraph  systems,  where  the  high 
conductor  cost  warrants  expensive  terminal  equipment  for  reasons  of  efficiency  and 
economy.  However,  for  land  line  teletypewriter  commercial  use,  neither  expensive  cir- 
cuits nor  elaborate  terminal  arrangements  are  Justified,  and  the  facilities  and  equipment 
that  are  provided  represent  a  compromise  on  a  cost  balance  basis. 

For  good  telegraph  transmission  with  comparatively  simple  equipment,  it  is  desirable 
that  the  received  signal  contain  a  substantial  part  of  the  second  and  third  harmonics  of 
the  frequency  of  the  unit  signal  element.  For  60-speed  teletypewriter  signals,  the  shortest 
signal  element  is  0.022  sec,  which  is  equivalent  to  1/2  cycle.  The  fundamental  frequency 
at  this  speed  then  would  be  I/ (0.022  X  2)  =  22.73  cycles  per  second,  the  third  harmonic  of 
which  would  be  about  68.2  cycles  per  second. 

If  one  teletypewriter  character  is  continuously  repeated  at  60-speed,  the  signal  wave 
(there  being  one  per  character)  repeats  itself  about  6.1  times  per  second,  so  that,  for  the 
purpose  of  discussing  the  signal  components  in  teletypewriter  operation,  the  wave  may  be 
considered  as  composed  of  a  d-c  component  and  harmonics  of  a  fundamental  frequency 
of  6.1  cycles,  rather  than  harmonics  of  the  signal  element  frequency.  Thus,  in  Fig.  11, 
the  left-hand  column  shows  several  of  the  more  important  harmonic  components  of  the 
D  signal,  their  relative  magnitudes  and  phase  relationships,  the  first  harmonic  (curve  6) 
being  shown  as  a  sine  wave  of  the  same  time  period  as  the  overall  signal;  the  right-hand 
column  shows  the  combined  resultant  of  the  d-c  component  and  the  overall  signal  har- 
monics, added  successively,  as  indicated. 

Since  the  fundamental  signal  element  frequency  for  60-speed  operation  was  shown  previ- 
ously to  be  22.7  cycles  per  second,  theoretically  the  character  D  could  be  interpreted  correctly 
by  an  ideal  receiving  device  if  components,  in  correct  phase  relation,  up  to  and  including 
the  fourth  harmonic  of  the  overall  signal  (curve  o  in  right-hand  column)  were  received. 

In  practice,  overall  signal  harmonics  up  to  about  the  tenth  (corresponding  to  about  the 
third  harmonic  of  the  signal  element  frequency)  are  transmitted,  which  gives  a  signal  wave 
similar  to  curve  r  and  which  resembles  somewhat  the  square-corner  wave  in  Fig.  11. 
*  CARRIER  TELEGRAPH.  Carrier  telegraph  systems  employ  modulated  alternating 
currents  of  different  frequency  bands.  The  voice-frequency  (low-frequency)  system  uti- 
lizes the  band  of  255  to  3145  cycles 
for  18  channels  of  170  cycles  per 
channel.  The  modulated  output  of 
this  system  may  be  impressed,  if 
desired,  on  broad-band  telephone 
carrier  or  radio  channels  of  higher- 
frequency  bands,  making  use  of  such 
channels  for  telegraph  rather  than 
for  telephone  service.  The  high- 
frequency  system  for  telegraph  was 
the  first  one  developed.  It  is  for 
open-wire  facilities  and  operates  in 
three  different  frequency  assign- 
ments, all  of  which  lie  within  the 
overall  range  of  3.33  kc  to  11.25  kc. 
These  systems  will  be  described  later 
in  this  section. 

As  previously  described,  the  basic 
signal  element  is  a  square  wave  pro- 
duced by  making  and  breaking  a  d-c 


A.  Impressed  telegraph  current 


B.  unmodulated  carrier  current 


0.  Carrier  after  modulation 


FIG.   12.     D-c  Modulation   of   Carrier   Currents — Tele- 
graph Carrier  Systems  (Courtesy  Bell  System) 


sending  circuit  or  by  applying  opposite  potentials  (one  at  a  time)  to  the  transmitting  de- 
vice. Generally,  the  carrier  wave  is  supplied  continuously  to  the  transmitting  device  and 
is  interrupted  (modulated)  in  accordance  with  the  d-c  signals  being  produced  by  the  local 
sending  circuit  and  impressed  on  the  carrier  wave  at  this  device.  Figure  12  shows  the 
impressed  telegraph  current  (polar  operation),  the  unmodulated  carrier  current,  and  the 


DISTORTION 


18-11 


modulated  carrier  wave.  The  received  segments  of  carrier  current  signal  are  rectified  in 
the  demodulator  circuit,  and  the  resulting  unidirectional  plate  current  pulses  act  on  a  re- 
ceiving d-c  relay.  These  received  d-c  current  signals,  being  similar  in  pattern  to  the  sent 
d-c  signals,  actuate  the  receiving  relay  in  accordance  with  the  sent  signal. 

Since  the  sending  d-c  signal  contains,  for  practical  reasons,  up  to  about  the  third  har- 
monic, which  at  60-speed  operation  is  about  70  cycles,  both  carrier  sidebands  transmitted 
will  also  contain  this  band  width,  or,  for  both  bands  at  this  speed,  the  frequency  spread 
would  be  about  140  cycles.  For  higher  or  lower  speeds,  this  spread  would  be  greater  or 
less. 

5.  DISTORTION 

An  ideal  (perfect)  telegraph  circuit  reproduces  telegraph  signals  at  the  receiving  end 
exactly  as  they  are  impressed  at  the  sending  end,  with  respect  to  length,  but  not  necessarily 
amplitude,  of  the  component  marks  and  spaces.  The  time  of  travel  of  the  signal  over  a 
circuit  is  usually  not  important,  even  though  such  time  exceeds  the  duration  of  a  unit 
signal  element. 

Distortion  in  telegraph  transmission  is  thus  determined  by  comparing  the  length  and 
relative  arrangement  of  the  signal  elements  as  sent  with  the  length  and  relative  arrange- 
ment of  such  elements  as  finally  delivered  by  the  receiving  device. 

The  overall  or  total  resultant  distortion  of  signals,  for  a  given  telegraph  circuit,  consists 
of  two  principal  types  of  distortion,  namely  systematic  and  fortuitous,  which  result  from 
a  number  of  different  causes  and  require  different  treatments  in  design  and  maintenance 
work  in  order  to  meet  service  requirements. 

Assume  that  a  given  telegraph  character  is  sent  continuously  over  a  telegraph  circuit 
and  that  each  repetition  of  the  character  is  considered  perfect  as  sent.  Measurements  of 
the  distortion  of  each  of  the  unit  marks  (elements)  in  a  large  number  of  successive  repeti- 
tions of  the  character  at  the  receiving  end  will  generally  indicate  that  the  distortion  differs 
(1)  from  element  to  element  in  a  given  repetition  of  the  character  and  (2)  from  character 
to  character  for  a  particular  element  in  the  character.  The  average  of  a  large  number  of 
distortions  for  a  particular  element  is  designated  as  systematic  distortion.  The  individual 
departure  of  the  distortion  from  the  average  for  a  given  measured  distortion  is  designated 
as  fortuitous  distortion.  The  total  distortion  of  each  signal  element  is  the  algebraic  sum 
of  the  systematic  and  fortuitous 

distortions  and  is  the  amount  of  Duration  of  sent  signal  elements  (=100  3D 

deviation  between  the  sent  and  ~" 

received  signals. 

Figure  13  illustrates  roughly 
for  10  repetitions  (not  enough  for 
a  good  average  in  actual  cases) 
the  distortion  which  may  affect  a 
marking  signal  of  unit  duration 
as  received.  The  average  length 
of  received  signal  is  shown  to  be 
90  per  cent  of  the  sent  signal, 
resulting  in  a  —  10  per  cent  sys- 
tematic distortion  which  applies 
to  all  the  repetitions.  The  de- 
parture of  the  individual  distor- 
tions from  the  average  varies 
between  repetitions,  as  shown  in 
the  right-hand  column,  and  in 
the  left-hand  column  the  result- 
ant distortion  is  indicated. 

Systematic  distortion  may  be 
divided  into  two  component  dis- 
tortions, bias  and  characteristic, 
for  the  purpose  of  analyzing  and 
treating  the  causes  of  this  type  of 
distortion.  The  nature  of  these 

components  may  be  explained  by  assuming  a  telegraph  system  in  which  marks  and  spaces 
are  sent  by  means  of  currents  equal  in  magnitude  but  of  either  positive  and  negative  or 
negative  and  positive  sign,  respectively,  as  desired. 

Bias  Distortion.  Assume  that  the  systematic  distortion,  as  measured,  is  due  to  a  higher 
positive  than  negative  sending-end  potential,  and  that  this  fact  results  in  lengthening 


Duration  of  received  sign  a  I  elements 

n-  *  -*•      Systematic  Fortuftoas 

Distortion  (exponent         component 


1st 

1 

-1 

0% 

-10% 

9 

2nd 

> 

J, 

41056 

Srd 

-li 

* 

-10* 

9 

4th 

*« 

* 

-1055 

42056 

•5th 

-2C 

)* 

->* 

4* 

4« 

* 

-J* 

430* 

TtTi 

-4C 

,< 

4* 

-ail 

8th 

-« 

* 

J* 

0 

Q>h 

-1 

y% 

j>* 

0 

10th 

-3C 

& 

j* 

-20* 

i         I 

FIG.  13. 


Average  distortion^  LO  %  ^systematic  com  ponent 

Systematic  and  Fortuitous  Distortion   (Courtesy 
Bell  System) 


18-12 


TELEGRAPHY 


the  marks  when  positive  current  is  used  for  transmitting  marks,  and  in  shortening  the 
marks  when  negative  current  is  used  for  transmitting  marks.  Then,  interchanging  the 
functions  of  the  two  current  conditions  employed  changes  the  sign  of  the  systematic 
distortion  but  not  its  magnitude  and  is  called  bias  distortion,  indicating  a  lack  of  sym- 
metry in  the  circuit. 

A  marking  bias  is  called  a  positive  bias,  and  a  spacing  bias  is  called  a  negative  bias;  and, 
since  the  lengths  of  the  marks  and  spaces  may  be  indicated  in  milliseconds  (ms) ,  the  amount 
of  bias  may  also  be  indicated  in  milliseconds  by  the  formula  (M-STD)  —  (S-MTD)  = 

ms  bias,  the  first  two  terms  being 
mark-to-space  transition  delay  and 
space-to-mark  transition  delay,  re- 
spectively, and  the  sign  of  the  result 
(using  this  formula)  being  the  sign  of 
the  bias.  Thus,  for  a  given  circuit, 
if  M-STD  =  6  ms  and  S-MTD  *  3 
ms,  the  milliseconds  bias  is  +3,  in- 
dicating that  every  mark,  regardless 
of  length,  will  be  increased  3  ms,  and 
every  space,  regardless  of  length, 
will  be  decreased  3  ms. 

Though  a  given  ms  bias  condition 


Current  In  mllllamperes 

i->  w  •£"•  o>  s 
O  Cn  O  tn  o  <- 

—S-MTD's 

•«$ 

p=V 

/"" 

V 

-M-STD's 

A- 

) 

\ 

; 

<  T0  \ 

d 

r 

T 

\ 

t> 

|\ 

/ 

/  ' 

. 

C 

V 

3                       10                     20 

30                     4( 

FIG.   14o. 


Time  in  milliseconds 

Effect   of  Relay  Biasing    Current   on   Signal 
Length  (Courtesy  Bell  System) 


is  due  to  such  factors  as  unequal 
marking  and  spacing  line  currents, 
ground  potentials,  and  improper 
biased  relay  adjustments,  this  con- 
dition will  remain  constant  for  the 
transmission  of  signal  reversals  (but  not  random  signals)  irrespective  of  the  speed  of 
transmission.  However,  the  effect  of  a  given  ms  bias  condition  on  transmission  does  vary 
with  the  length  of  the  transmitted  marks  and  spaces.  For  example,  in  a  manual  telegraph 
circuit  the  dashes  (long  marks)  are  about  3  times  the  length  of  the  dots  (short  marks), 
and  both  dashes  and  dots  decrease  proportionately  in  length  with  increase  in  speed  of 
transmission.  Assuming  first  a  slow  speed,  where  the  dots  are  30  ms  long  and  the  dashes 
are  90  ms  long,  a  ms  bias  of  -f-10  will  lengthen  the  dots  and  dashes  to  40  ms  and  100  ms, 
respectively,  the  ratio  of  dash  to  dot  length  still  being  2.5  to  1.  However,  if,  owing  to 
an  increase  in  speed,  the  dots  and  dashes  are  shortened  to  5  ms  and  15  ms,  respectively, 
the  same  +10  ms  bias  would  result  in  dots  15  ms  in  length  and  dashes  25  ms  in  length. 
The  ratio  of  dash-to-dot  length  would  then  be  about  1.7  to  1,  and  greater  difficulty  would 
be  experienced  in  reading  the  signals  than  for  the  usual  ratio  of  2.5  or  3  to  1. 

Figures  14a  and  b  show  the  effects  on  signal  length  of  relay  biasing  current  and  line  cur- 
rent variations.  In  the  first  figure, 
the  line  current  is  held  constant  at 
60  ma,  while  the  normal  biasing  cur- 
rent (line  B)  is  increased  (lineiA)  and 
decreased  (line  C).  In  the  second 
figure,  the  biasing  current  is  held  con- 
stant and  the  normal  line  current  is 
increased  (high  line  current)  and  de- 
creased (low  line  current) .  It  is  evi- 
dent that,  in  the  first  condition,  in- 
creasing the  biasing  current  decreases 
the  signal  length  and  decreasing  it 
increases  the  signal  length;  and,  in 
the  second  condition,  increasing  the 
line  current  increases  the  signal  length 
and  decreasing  it  decreases  the  signal 
length. 

Characteristic  distortion  results 
from  various  causes,  which  are  usually  different  from  those  associated  with  bias  distortion. 
Assume  a  telegraph  system,  in  which  the  sending  battery  potentials  are  equal  and  oppo- 
site in  sign  and  in  which  the  marks  and  spaces  are  formed  by  corresponding  currents, 
equal  but  opposite  in  sign.  Also,  assume  that,  owing  to  the  characteristics  of  the  given 
system,  the  current  is  slow  in  building  up  to  the  normal  mark  or  space  value.  If  the  cur- 
rent does  not  have  time  to  reach  its  final  value  on  the  short  signal  elements,  the  first 
mark  following  a  long  space  may  be  shortened.  Under  this  condition,  it  is  obvious  that 
interchanging  the  functions  of  the  positive  and  negative  current  will  not  alter  either  the 


330 


20 
Time  in  milliseconds 


FIG.  14&.     Effect  of  Line  Current  Magnitude  on  Signal 
Length  (Courtesy  Bell  System) 


DISTORTION 


18-13 


sign  or  the  magnitude  of  the  resulting  distortion,  and  the  distortion  is  called  characteristic 

distortion,  indicating  it  is  a  function  of  the  signal  combination  and  fixed  characteristics 

oi  tne  system,  which  causes 

remnants  from  a  given  signal 

element  to  affect  succeeding 

elements. 

Depending  on  the  speed  of 
operation,  telegraph  signals 
are  frequently  of  insufficient 
duration  to  permit  the  line 
current  for  a  given  signal  to 
change  from  one  steady  state 
to  the  other,  i.e.,  from  a  maxi- 
mum positive  to  a  maximum 
negative  line  current,  or  vice 
versa.  In  such  cases,  the 
transition  M-S  or  S-M  will 
occur  while  the  current  is 
changing,  and  this  is  desig- 
nated as  a  changing-current 
M-S  or  S-M  transition,  both 
of  which  are  shown  in  Fig.  15. 
It  will  be  noted  that,  for  this 
particular  polar  telegraph 
system,  the  time  for  the  line 


1 

k, 

o 


ao 

§    20 

1  10 


30  40 

Time  in  milliseconds 

A 


£-10 


-30 


—40 


"1   J 

Steady-state 
x1  —  M-S  transition 

^  

\ 

i 
<  —  20  ms- 

> 

/ 

\ 

S-MTD 

Q     m<; 

!\ 

/ 

f 

M-STD, 

\ 

/ 

10 

ms' 

N 

\ 

/ 

'Xs^    Changing  current 
^-S-M  transition 

N, 

/ 

33  ms— 

"*"*-—  4 

10             20             30             40            50             60           70 
Tlrjie  in  milliseconds' 

B 

current  to  change  from  one 
steady-state  condition  to  the 
other  is  33  ms,  whereas  the 
duration  of  the  marking  or 
spacing  signal  is  only  22  ms. 
The  net  effect  on  the  signal 
is  to  shorten  it  2  ms,  since  the 
transition  delay  at  the  start 
of  the  signal  is '10  ms  (sub- 
tracts from  the  signal  length) 
and  at  the  end  of  the  signal  is 
only  8  ms  (adds  to  the  signal 
length).  The  total  current  change  in  either  transition  is  28  ma  (+25  to  —3  or  —25  to 
+3) ;  but,  if  the  transmitted  signal  had  been  longer  than  22  ms,  the  line  current  would 
have  exceeded  plus  or  minus  25  ma  and  at  the  end  of  the  signal  the  transition  delay 


FIG.  15.     Changing  Current  Transitions  (Courtesy  Bell  System) 


*• 
O 


30  40 

Time  in  milliseconds 


FIG.  16.    Characteristic  Distortion  Effects  on  Signal  Lengths  at  40f  60,  and  75  Speed  Operation  (Cour- 
tesy Bell  System) 

would  have  been  greater,  up  to  the  limiting  delay,  resulting  from  the  line  current  in- 
creasing, up  to  its  steady-state  value. 

Also,  if  the  transmitted  signal  had  been  less  than  22  ms  long,  the  line  current  -would  not 
have  built  up  to  25  ma  by  the  time  the  signal  ended,  and  the  transition  delay  at  the  end 


18-14 


TELEGRAPHY 


of  the  signal  would  have  been  less  than  for  22  ms.  Figure  16  shows  characteristic  distor- 
tion effects  on  marking  signal  length  at  40,  60,  and  75  teletypewriter  operating  speeds. 
The  time  required  for  the  line  current  to  change  from  its  maximum  negative  to  positive 
value,  and  vice  versa,  is  33  ms,  and  the  marking  signals  are  33  ms,  22  ms,  and  18  ms  long, 
respectively.  Changing  current  transitions  take  place  for  all  except  the  33  ms  signal 
(40  speed),  for  which  the  steady-state  current  values  are  just  attained.  It  will  be  noted 
that  the  S-MTD  values  are  the  same  (10  ms)  for  all  signal  lengths,  but  the  M-STD  values 
decrease  with  decreasing  signal  length.  For  similar  spacing  signal  lengths,  the  M-STD 
values  would  be  the  same  (10  ms),  while  the  S-MTD  values  would  decrease  with  decreas- 
ing signal  length. 

The  amount  of  a  changing  current  transition  delay  is  thus  dependent  on  the  line  current 
value  at  the  start  of  the  transition,  and  this  line  current  value  depends  on  the  signal  length 
or  speed  of  transmission.  The  lengths  of  the  received  signals  are  obviously  affected  by 
these  changing  current  transitions,  and  the  magnitude  of  the  effect  is  inversely  propor- 
tional to  the  length  of  the  sent  signals.  Since  received  short  signals  are  shortened  by  this 
effect,  the  distortion  is  known  as  negative  characteristic  distortion.  An  opposite  effect  is 
possible,  though  not  common,  and  may  result,  if  the  line  current  increases  momentarily 
at  the  end  of  each  transition  to  a  value  exceeding  the  steady  state,  owing  to  circuit  charac- 
teristics and  transient  effects.  This  effect  would  tend  to  lengthen  the  received  mark  or 
space  signal  and  would  be  known  as  positive  characteristic  distortion. 

Some  of  the  principal  differences  between  bias  and  characteristic  distortion  are  given  in 
Table  I. 

Table  1.    Differences  between  Bias  and  Characteristic  Distortion 


Type  of  Distortion  Is  Affected  by 

Type  of  Distortion 

Bias,  ms 

Characteristic 

1  .  Length  of  signal  

No 
Yes 

No,  except  in  neu- 
tral operation 
Yes 

No 
Yes 

Yes 

No 

Yes 

No 

Yes 
Not  appreciably 

2.  For  a  given  length  of  signal,  whether  the  signal  is  marking 
or  spacing. 
3.  Amount  and  arrangement  of  the  circuit  capacitance,  induct- 
ance, and  resistance. 
4.  Unequal  marking  and  spacing  line  currents. 
Change  in  line  current. 
Change  in  receiving  relay  biasing  current. 
Ground  potential  difference  between  sending  and  receiving 
end. 
5,  Speed  of  transmission 

6.  Usual  operating  variations,  occurring  frequently  throughout 
the  day,  such  as  voltage  fluctuations  and  relays  requiring 
adjustments. 

Measurements  of  systematic  distortion,  in  practice,  will  usually  indicate  that  both  bias 
and  characteristic  distortion  are  present,  or  the  total  measured  distortion  with  the  circuit 
normal  is: 

SI  (Total  distortion,  circuit  normal)  =  C  (characteristic)  +  B  (bias) 

and  the  total  measured  distortion  with  the  reversed  condition  is: 

S2  (Total  distortion,  circuit  reversed)  =  C  (characteristic)  —  B  (bias) 

The  characteristic  component  is  (Si  +  S^)  /2,  and  the  bias  component  is  (Si  —  Sz)  /2. 

Referring  to  Fig.  17,  assume  that  a  repeated  signal  is  being  sent  over  a  given  circuit, 
consisting  of  a  marking  element  1  unit  long  and  a  spacing  element  3  units  long.  If  no 
distortion  exists,  this  signal  will  be  received  exactly  as  sent.  However,  when  measured 
with  the  circuit  normal,  the  unit  mark  is  found  to  be  15  per  cent  too  long,  as  shown  at  A. 
If  a  measurement  is  now  taken  with  the  line  conditions  for  marking  and  spacing  reversed, 
this  unit  mark  is  found  to  be  5  per  cent  too  long,  as  shown  at  B.  By  formula,  the  char- 
acteristic distortion  C  is  10  per  cent,  and  the  bias  distortion  B  is  5  per  cent,  both  with 
signs  positive  or  marking.  If,  with  the  line  conditions  reversed,  the  unit  mark  was  found 
to  be  5  per  cent  too  short,  as  shown  at  £',  the  characteristic  component  of  the  15  per  cent 
marking  distortion  (shown  at  A)  would  be  5  per  cent  and  the  bias  component  10  per  cent, 
both  positive  or  marking. 

In  practice,  systematic  distortion  is  usually  determined,  with  the  same  methods  as  above, 
by  measuring  a  repeated  signal  consisting  of  a  short  marking  signal  followed  by  a  long 
spacing  signal  for  the  normal  condition  and  then  a  repeated  signal  consisting  of  a  short 
spacing  signal  followed  by  a  long  marking  signal  for  the  reversed  condition. 


DISTORTION 


18-15 


Una 
•6  —  signal  —  > 
element  —  $. 

•"-  

u- 

_j  Cl 

U-5* 

}    com 

id 

*=£% 

*^cond 

B 


istortion  with  circuit 
^CjwpTti pjis_intercha  nged=Sg 


distortion  with  circuit    f 
iditions  intsrchaiTged=.S2 


For  A  and  B: 
Characteristic  distortion 


For  A  and  B: 
Characteristic  Distortion 


BIas= 
FIG.   17. 


Though  the  sign  of  the  final  value  of  characteristic  distortion,  as  computed,  is  not 
important,  the  sign  for  bias  distortion  is  always  important  and  is  indicated. 

Fortuitious  distortion  is  caused  by  such  factors  as  crossfire,  power  induction,  momentary 
battery  fluctuations,  line  hits,  break  key  operation,  and  similar  effects.  This  type  of  dis- 
tortion acts  to  alter  the  received  signals  by  various  amounts  in  an  irregular  manner.  In 
transmitting  miscellaneous  signals,  the  combined  effect  of  all  distortion  on  the  displacement 
of  received  transitions  may  result  in 

signals,  sometimes   called  "jitter,"        \* Total  length  of  signal 

because  of  their  rapid  variations,  or 
the  effect  may  cause  a  complete 
breakdown  in  signal  transmission. 

Total  distortion,  for  any  given  tele- 
graph circuit,  is  usually  a  combination 
of  its  bias,  characteristic,  and  fortu- 
itous components,  and  the  total  distor- 
tion determines  the  quality  of  telegraph 
transmission.  However,  for  reasons 
of  design  and  maintenance,  it  is  usu- 
ally desirable  to  determine  also  the 
value  of  these  components. 

Telegraph  distortion  is  usually 
given  as  a  percentage  of  a  perfect  sig- 
nal element  of  unit  length.  For  man- 
ual operation,  these  elements  or  dots 
are  usually  sent  at  a  rate  of  12  or  13 
dots  per  second,  and  then*  duration 
(taking  into  account  the  dot  length 
interval  between  each  two  dots)  is 
about  40  ms.  For  60-speed  tele- 
typewriter transmission,  the  signal- 
ing rate  is  equivalent  to  about  22,7 
dots  per  second,  and  the  duration  of 
the  unit  signal  element,  in  this  case, 
is  22  ms. 

TELETYPEWRITER  DISTORTION.  Distortion  as  it  affects  start-stop  teletypewriter 
signals  must  be  considered  in  a  somewhat  different  manner  from  that  for  other  types  of 
telegraph  systems,  because  teletypewriter  system  operation  differs  fundamentally  from 
the  operation  of  other  systems.  Each  teletypewriter  operation  (the  transmission  of  a 
character),  as  previously  described  in  this  section,  is  initiated  by  the  mark-to-space  transi- 
tion at  the  beginning  of  the  start  pulse  of  the  received  character.  It  is  important  that 
each  of  the  succeeding  transitions  in  the  character  be  correctly  timed  with  respect  to  the  first 
transition.  Distortion  causes  the  displacing  of  succeeding  transitions  from  their  normal 
positions  with  respect  to  the  start  transition  and  thus  reduces  the  margin  of  operation  of 
the  teletypewriter. 

Bias  distortion  usually  affects  both  the  beginnings  and  ends  of  the  received  signal 
elements.  However,  the  teletypewriter  receiving  mechanism  starts  on  a  mark-to-space 
transition  (for  each  character  received),  and  this  transition  is  also  affected  by  the  same 
bias,  so  that  succeeding  mark-to-space  transitions  in  the  character  will  not  be  displaced 
with  respect  to  the  start  transition,  owing  to  bias.  The  space-to-mark  transitions  will  be 
displaced  with  respect  to  the  start  transition  by  an  amount  equal  to  the  total  bias.  With 
marking  bias,  all  space-to-mark  transitions  will  be  uniformly  displaced  toward  the  start 
transition,  whereas  with  spacing  bias  they  will  be  uniformly  displaced  away  from  the  start 
transition. 

Characteristic  distortion  may  displace  both  the  received  space-to-mark  and  mark-to- 
space  transitions  with  respect  to  the  start  transition,  depending  on  the  signal  combina- 
tion, and  recurs  for  the  same  signal  combination.  It  may  affect  both  ends  of  the  teletype- 
writer orientation  range,  and  when  miscellaneous  characters  are  received  a  distinction 
cannot  be  made  between  the  characteristic  and  fortuitous  components  of  the  distortion. 

Fortuitious  distortion  displaces  miscellaneous  received  transitions  by  various  amounts 
in  a  random  manner,  regardless  of  the  signal  combination.  These  effects  may  result  in 
errors  in  received  characters  or,  if  severe,  in  complete  circuit  failure. 

The  total  distortion  is  the  displacement  of  a  received  transition  from  its  correct  time  of 
occurrence,  and  it  is  equal  to  the  algebraic  sum  of  the  fortuitous  distortion  and  the  systematic 
distortion.  The  total  distortion  determines  the  margin  of  operation  of  the  receiving  tele- 
typewriter, and  it  is  a  measure  of  the  transmission  quality  of  the  received  signals 


Characteristic  and  Bias  Distortion 
Bell  System) 


(Courtesy 


18-16 


TELEGRAPHY 


In  teletypewriter  operation,  not  only  signal  distortion  takes  place  during  transmission, 
but  also  mechanical  variations  occur  in  the  teletypewriter  mechanisms  which  affect  the  qual- 
ity of  signal  transmission.  The  mechanical  operation  of  the  teletypewriter  is  described 
briefly  in  article  7  of  this  section,  but,  for  the  purpose  of  discussing  distortion  as  it  affects 
teletypewriter  operation,  some  of  the  mechanical  features  of  the  teletypewriter  must  be 
referred  to  in  this  article. 

Figure  18  shows  some  of  the  principal  mechanical  units  of  the  selecting  arrangement  of 
a  teletypewriter  which  determines  the  instant,  in  the  duration  of  a  given  received  signal 
element,  when  the  actual  selection  takes  place.  For  example,  if  the  signal  element  being 
received  is  marking,  the  selection  must  be  made  at  some  instant  during  the  time  this 


Acnratore  extensfoi 


.ocktog  fevar 


;k!hg  cam 


larking  bias 

Spacing  bias- 
Distortions  other 
than  bias 

Send  dist.  6.7%  faster 

than  rec.  dist. 
a  Send  dist.  5.9.%  slovyec 
*       than  rec.  dist. 

FIG.  18.     Distortion  Effects  on  the  Selection  of  Teletypewriter  Signals  (Courtesy  Bell  System) 

particular  element  is  being  received,  in  order  that  the  code  bar,  corresponding  to  this 
selection,  will  be  properly  positioned  and  the  character  of  which  this  element  is  a  part 
will  be  correctly  recorded. 

The  selecting  mechanism  includes:  (1)  the  line  magnet,  which  is  operated  by  marks 
and  non-operated  by  spaces;  (2)  the  armature  and  armature  extension,  which  are  held  in 
the  position  shown  during  marking  and  are  released  to  the  dotted  position  during  spacing; 
(3)  the  stop  latch,  which  stops  the  locking  cam  rotation  after  each  revolution;  (4)  the 
locking  lever,  whose  point  B,  during  rotation  of  the  locking  cam,  engages  one  side  or  the 
other  of  the  end  of  the  armature  extension,  depending  on  the  position  of  the  extension, 
thereby  locking  the  extension  momentarily,  which  results  in  the  code  bars  being  properly 
positioned;  and  (5)  the  locking  cam,  which  is  driven  by  a  friction  clutch  and  which,  in  a 
single  revolution,  causes  the  locking  lever  to  be  positioned  five  times,  once  for  each  of  the 
five  signal  elements  received,  during  the  transmission  of  one  character. 

In  the  idle  condition,  the  received  signal  is  always  marking  and  the  magnet  is  energized, 
but  the  selector  cam  assembly  is  held  stationary  by  the  stop  latch  until  the  start  pulse 
(always  spacing)  is  received,  when  the  stop  latch  is  released  and  the  cam  starts  rotating. 
The  speed  of  rotation  and  the  starting  position  of  the  selector  cam  assembly  are  normally 
so  adjusted  that  the  first  depression  on  the  locking  cam  (point  A)  will  arrive  at  the  locking 
lever  point  (which  rides  the  cam)  at  about  the  instant  the  middle  of  the  first  selecting  signal 
element  is  being  received.  At  this  instant  the  locking  lever  point  (riding  the  cam)  moves 
into  the  cam  depression,  and  the  lever  point  B  moves  forward  to  lock  the  armature  and 
its  extension  in  one  of  the  two  positions  they  may  occupy  at  that  instant.  The  position 
of  the  armature  extension  determines  which  of  the  two  line  conditions,  marking  or  spac- 
ing, will  be  recorded  for  the  signal  element  being  received,  and  the  position  of  the  cor- 
responding code  bar.  As  the  cam  rotates,  a  similar  selection  takes  place  as  the  locking 
lever  enters  each  of  the  five  depressions  on  the  cam  and  the  corresponding  code  bars  are 
positioned  in  succession.  The  final  arrangement  of  the  five  code  bars  results  in  the  selec- 


DISTORTION  18-17 

tion  of  one  and  one  only  type  bar,  which,  when  actuated  immediately  after  completion  of 
one  revolution  of  the  locking  cam,  prints  the  sent  character  on  paper.  The  final  pulse  of 
the  train  of  signal  elements  (for  each  character  received)  is  always  marking  and  somewhat 
longer  (1.42  units)  than  the  preceding  six  elements  (1.0  unit  each).  During  this  pulse,  the 
stop  arm  on  the  receiving  selector  cam  assembly  strikes  the  stop  latch,  and  the  assembly 
is  then  held  stationary  until  the  next  start  pulse  is  received  for  the  next  character. 

When  a  start  pulse  is  received,  a  small  increment  of  time  will  elapse  before  the  selector 
cam  assembly  attains  full  speed,  owing  to  such  factors  as  the  inertia  of  moving  parts  and 
clutch  slippage.  This  delay  is  compensated  for  by  slightly  decreasing  the  distance  between 
the  point  at  which  the  locking  lever  rests  on  the  locking  cam,  and  point  A  on  the  cam,  from 
what  the  distance  would  be  if  these  factors  were  not  present.  This  adjustment  is  repre- 
sented by  the  distance  x,  as  shown  on  the  cam  surface  in  Fig.  18. 

The  teletypewriter  is  usually  equipped  with  an  orientation  or  range  finder  device,  which 
permits  rotating  the  stop  latch  with  respect  to  the  locking  lever,  thus  changing  the  time 
of  selection  with  respect  to  the  start  signal.  The  range  finder  moves  the  stop  position 
either  forward  or  backward  and  has  a  pointer  which  moves  along  a  scale,  the  scale  being 
calibrated  in  percentage  (0  to  120)  of  a  unit  signal  element.  If  the  pointer  is  moved  toward 
the  lower-numbered  part  of  the  scale,  the  time  between  the  start  and  selecting  points  is  reduced 
and  the  time  of  selection  is  advanced  toward  the  beginning  of  each  selecting  element. 
//  the  pointer  is  moved  toward  the  higher-numbered  part  of  the  scale,  the  time  between  the  start 
and  selection  is  lengthened  and  the  time  of  selection  is  moved  toward  the  end  of  each  select- 
ing element.  For  an  ideal  teletypewriter,  whose  mechanism  acted  instantly  and  selected 
exactly  the  corresponding  instant  of  each  signal  element,  the  range  finder  could  be  moved 
over  a  range  of  100  per  cent,  if  perfect  signals  were  received,  without  causing  errors  in  the 
received  signals,  as  shown  by  (a)  in  Fig.  IS.  Moving  the  range  finder,  in  effect,  shifts  the 
solid  vertical  lines  with  respect  to  the  signal  elements,  and  for  (a)  the  time  of  selection 
could  be  changed,  without  error,  by  ±50  per  cent.  Practically,  teletypewriter  specifica- 
tions require  the  overall  range  to  be  at  least  72  per  cent  without  error  for  perfect  signals. 

Distortion  in  teletypewriter  signals,  as  previously  stated,  is  usually  some  combination 
of  bias,  characteristic,  and  fortuitous  distortion.  Theoretically,  if  bias  exceeds  50  per  cent 
in  (c)  and  (e)  of  Fig.  18T  errors  will  result  in  the  recorded  signal,  but,  from  a  practical 
standpoint,  the  bias  tolerance  is  of  the  order  of  ±40  per  cent  with  perfect  received  signals 
because  of  allowances  that  must  be  made  for  other  variations. 

Internal  bias  may  exist  in  a  teletypewriter  as  the  result  of  such  factors  as  improper 
adjustment  of  the  line  relay  or  receiving  magnet.  This  bias  reduces  the  orientation  margin 
more  when  receiving  perfect  signals  than  when  receiving  signals  with  a  bias  equal  in 
magnitude  but  opposite  in  sign.  In  order  to  minimize  tin's  bias  effect,  the  range  finder 
should  be  set  at  the  point  where  signals  having  equal  marking  and  spacing  bias  just  cause 
errors. 

The  effect  of  shortening  the  start  pulse  by  25  per  cent,  (f)  in  Fig.  18,  is  equivalent  to 
lengthening  the  stop  pulse  and  retarding  the  points  of  selection  for  the  succeeding  signal 
elements  by  the  same  amount.  The  effect  of  shortening  the  end  of  the  stop  pulse  by  25 
per  cent  (g)  is  equivalent  to  advancing  the  points  of  selection  for  the  succeeding  signal 
elements  by  the  same  amount.  The  effect  of  advancing  the  beginning  of  element  1  and 
of  retarding  the  beginning  of  element  3  and  the  ending  of  element  4,  as  shown  by  (&),  will 
not  result  in  errors  in  the  received  character  if  the  range  finder  is  set  at  its  midpoint.  The 
effect  of  speed  variations  where  the  sending  machine  is  faster  than  the  receiving  machine  is 
illustrated  by  (i)»  and  where  the  reverse  is  true  is  shown  by  0")-  The  most  probable  error, 
in  both  cases,  would  be  in  the  proper  selection  of  element  5,  since,  in  the  first  condition,, 
part  of  the  stop  pulse  is  received  on  the  5  position,  and,  in  the  second  condition,  either 
element  4  is  extended  into  position  5  or  element  5  is  so  delayed  in  starting  that  it  would 
not  be  properly  received  in  its  normal  position.  The  effect  on  teletypewriter  margin  of 
speed  variation  is  mostly  at  one  end  or  the  other  of  the  orientation  range,  depending  on 
the  relative  speeds  of  the  sending  and  receiving  machines.  Though  the  speed  differences 
in  Fig.  18  are  shown  large  for  purposes  of  illustration,  actually  these  differences  are  nor- 
mally about  1/2  per  cent  or  less. 

The  orientation  range  limits  are  determined  by  the  various  distortions  present  for  a 
given  teletypewriter  circuit.  It  is  not  possible  to  judge  these  distortions  quantitatively 
by  the  limits  obtained  when  more  than  one  type  of  distortion  exists.  However,  assuming 
low  machine  bias,  these  limits  do  give  a  good  indication  of  the  quality  of  the  received 
signals.  If  the  limits  are  reasonably  definite,  some  fixed  distortion,  such  as  bias  or  speed 
difference,  is  generally  present,  while,  if  there  is  a  certain  range  at  each  limit  over  which 
certain  characters  are  consistently  in  error,  characteristic  distortion  is  indicated.  If  there 
is  a  range  over  which  errors  occur,  but  not  consistently  on  certain  characters,  fortuitous 
distortion  is  most  likely  present. 


18-18 


TELEGBAPHY 


TELEGRAPH  SYSTEMS 

By  John  D.  Taylor 

Telegraph  systems  employ  various  types  of  telegraph  equipment  and  connecting 
mediums  and  various  methods  of  transmission.  Direct-current  telegraph  systems  are 
used  extensively  in  land  wire  operation  and  in  conjunction  with  radio  channels,  while 
low-  and  high-frequency  carrier  telegraph  systems  utilize  both  land  wire  lines  and  radio 
channels  as  mediums  of  transmission.  Radio  telegraph  systems  make  use  of  radio  channels 
between  radio  transmitting  and  receiving  equipments  and  usually  wire  line  extensions 
between  such  equipments  and  the  telegraph  circuit  terminals. 


6.  DIRECT-CURRENT  SYSTEMS 


Open-wire  telegraph  channels  are  generally  obtained  by  using  bare  telegraph  wires. 
They  may  also  be  provided  by  simplexing  or  compositing  open-wire  telephone  circuits,  as 

shown  in  Figs.  1  and  2, 

Station  A  Station  B 

Simplex  coll 


Telephone  line  circuit 


Sounder 


respectively. 

The  simplex  arrange- 
ment is  shown  in  Fig.  1. 
The  arrows  represent  the 
telegraph  line  currents, 
which  divide  equally  at 
the  junction  of  the  two 
halves  of  the  line  wind- 
ings of  the  repeating  coil 
at  A  and  travel  over  both 
line  wires  of  the  telephone 
circuit  to  the  repeating 
coil  at  B,  where  they 
again  combine  at  the 

«.      i      j  ^ -,    ,         ^.      ..  ,^      x        junction  of  the  two  halves 
5n  Simplexed  Telephone  Circuit  (Courtesy    *f  ,-,  •         •-.  -i..,.     & 
Bell  System)  °*  ™s  COL^  De*ore  passing 

to    Station    B    telegraph 

equipment.  This  equal  division  of  the  current  is  possible  only  if  the  two  halves  of  the  line 
side  of  each  coil  are  identical  electrically  and  the  two  line  wires  have  identical  electrical 
characteristics.  To  the  extent  that  these  conditions  are  not  met,  the  current  will  not 
divide  or  combine  equally  at  the  coils  and  a  residual  induced  current  will  flow  in  the  drop 
windings  of  the  repeating  coils,  causing  interference  (Morse  thump)  in  the  telephone  cir- 
cuit. In  practice,  Morse  thump  becomes  objectionable  only  when  faults  occur. 


_       ,     _  .         ,    _. 
FIG.  1.    Telegraph  Circuit 


Telegraph 


.Retardation  colls 


Telegraph  leg 


Line  T-TJT 

wire         4rXjrl: 


FIG.  2.     Telegraph  Circuit  on  Composited  Telephone  Circuit  (Courtesy  Bell  System) 

The  composite  arrangement  is  shown  in  Fig.  2.  One  grounded  telegraph  channel  is 
obtained  from  each  line  wire  of  the  telephone  circuit  by  connecting  composite  sets  at  each 
terminal  of  the  circuit  in  such  a  manner  as  to  maintain  the  balance  of  the  circuit  and 


DIRECT-CURRENT  SYSTEMS 


18-19 


avoid  excessive  telegraph,  current  flow  in  the 
line  and  drop  windings  of  the  repeating  coils, 
which  current  would  cause  Morse  thump.  The 
equipment  on  the  two  sides  of  the  circuit  must 
be  well  balanced  and  the  line  wires  must  be 
closely  identical  electrically  to  prevent  tele- 
graph signals  from  interfering  with  the  tele- 
phone service.  With  this  arrangement  each 
telegraph  channel  is  independent  of  the  other 
and  is  usually  so  operated. 

The  composite  set  has  a  retard  coil  in  series 
with  the  telegraph  leg  and  capacitance  between 
the  leg  and  ground  to  prevent  sudden  changes 
in  signal  current  value  being  impressed  on  the 
line  wires  and  causing  current  surges  (clicks)  in 
the  telephone  circuit.  Also,  the  series  condens- 
ers in  the  telephone  circuit  prevent  direct  cur- 
rent from  reaching  the  repeating  coils  and 
assist  in  reducing  clicks,  while  the  retard  coil- 
condenser  bridge  on  the  drop  side  of  the  series 
condensers  functions  to  prevent  crossfire,  a  con- 
dition where  the  telegraph  signals  on  one  wire 
of  the  circuit  induce  potentials  on  the  other  wire 
that  interfere  with  the  telegraph  signals  over  it. 

A  schematic  diagram  of  a  neutral  telegraph 
circuit  is  shown  in  one  form  in  Fig.  6a,  article  4. 
This  circuit  operates  with  current  flowing  in 
either  direction  for  the  marking  condition  and 
no  current  flowing  for  the  spacing  condition. 
Both  sending  and  receiving  relays  usually 
operate  local  sounder  circuits,  which  produce 
the  recognized  audible  dots  and  dashes  of  the 
Morse  or  Continental  telegraph  codes  by  the 
armature  of  the  sounder  striking  its  front  and 
back  contacts,  corresponding  to  the  marking 
and  spacing  conditions.  Neutral  telegraph 
operation  is  also  employed  in  teletypewriter 
and  teleprinter  service  on  many  of  the  shorter 
circuits. 

The  receiving  station  may  break  the  circuit 
(stop  the  transmission  of  signals)  by  opening 
the  sending  key,  which  silences  both  sending 
and  receiving  sounders  and  indicates  to  the 
sending  station  that  the  receiving  station  de- 
sires to  send  signals. 

A  number  of  intermediate  stations  may  be 
connected  in  series  in  the  single  wire  line,  the 
number  depending  on  the  sensitivity  of  the  line 
relays,  the  battery  voltages  applied,  and  line 
conditions,  such  as  resistance  and  leakage. 

The  single  line  repeater,  a  schematic  dia- 
gram of  which  is  shown  in  Fig.  3,  may  be 
employed  over  long  single  wire  circuits.  This 
repeater  functions  to  receive  weak  signals  from 
the  line  from  either  direction  and  repeat  them 
with  normal  voltage  to  the  line  in  the  other 
direction.  The  signal  transmission  circuit 
through  the  repeater  is  shown  by  heavy  lines. 
Operation  of  the  west  station  key  causes  relay '  A 
to  repeat  signals  to  the  east  circuit,  and  operation 
of  the  east  station  key  causes  relay  A'  to  repeat 
signals  to  the  west  circuit. 

Opening  either  station  key  interrupts  current 
flow  in  both  east  and  west  lines,  and,  unless  aux- 
iliary circuits  were  provided  in  the  repeater, 


18-20 


TELEGRAPHY 


both  circuits  would  open  and  remain  open,  and  the  system  would  be  inoperative.  To 
prevent  this  condition,  a  biasing  and  a  locking  circuit  are  provided  (see  Fig.  3)  for  each 
direction  of  transmission.  Each  biasing  circuit  operates  the  armatures  of  the  line  and 
control  relays,  with  which  it  is  associated,  to  spacing,  when  the  circuit  through  the  line 
windings  of  these  relays  is  opened  during  the  transmission  of  signals.  However,  the  lock- 
ing circuit  is  so  designed  that,  for  east-to-west  transmission  with  the  east  circuit  open,  the 
biasing  current  reverses  direction  in  the  biasing  windings  of  the  line  west  relays  and  the 
armatures  of  these  relays  are  held  on  marking,  thus  maintaining  the  east  circuit  closed  at 
the  repeater,  regardless  of  the  position  of  the  east  station  key.  The  locking  circuit  func- 
tions oppositely  for  west-to-east  transmission. 

A  break  feature  (see  Fig.  3)  is  provided,  so  that  the  receiving  station  may  interrupt  the 
sending  station,  as  desired,  by  opening  the  receiving-station  key. 

The  one-way  polar  telegraph  circuit,  shown  in  Fig.  9,  article  4,  usually  employs  — 130 
volts  for  marking  and  +130  volts  for  spacing  signals  to  a  distant  polar  receiving  relay. 
This  circuit  operates  from  the  central  office  to  an  outlying  point  and  is  used  in  certain 
cases  where  a  one-way  service  only  is  required,  such  as  in  the  transmission  of  news  copy. 

Two -path  polar  operation  consists  essentially  of  two  one-way  polar  circuits  operating  in 
opposite  directions. 

Polarential  operation  provides  for  true  polar  operation  from  the  central  office  to  an 
outlying  point  and  a  modified  polar  operation  in  the  opposite  direction.  The  advantages 
of  polar  over  neutral  operation  (particularly  self-compensation  of  line  leakage)  are  thus 
obtained  with  relatively  simple  equipment  arrangements  at  the  outlying  point. 

Figure  4  shows  a  type  A  polarential  telegraph  circuit,  in  which  true  polar  signals  are 
transmitted  from  the  central  office,  and  ground  and  —130  volt  battery  (in  series  with  a 


i*— 1 

^+130  V.4 


Central  office 


FIG.  4.     Type  A  Polarential  Telegraph  Circuit  (Courtesy  Bell  System) 

total  added  resistance  of  990  ohms)  are  applied  in  transmitting  marks  and  spaces,  respec- 
tively, from  the  outlying  point.  In  this  case  a  repeater  is  not  used  at  the  outlying  point, 
as  is  more  commonly  done  in  type  A  operation. 

During  transmission  from  the  central  office,  the  circuit  is  closed  to  a  direct  ground  at 
the  outlying  point  through  the  sending  contacts  (in  this  case,  the  contacts  of  a  teletype- 
writer) while  the  central-office  receiving  relay  is  held  on  its  marking  contact  by  a  marking 
biasing  current.  This  current  is  adjusted  to  one-half  the  effective  spacing  current  when 
a  spacing  signal  is  being  sent  from  the  outlying  point.  During  transmission  from  the  out- 
lying point,  the  ground,  applied  for  a  marking  signal,  has  no  effect  on  the  central-office 
receiving  relay,  provided  that  the  line  and  artificial  line  are  balanced  at  the  duplex  set, 
and  this  relay  is  held  on  its  marking  contact,  while  negative  battery,  applied  through  the 
resistance  for  a  spacing  signal,  produces  an  effective  spacing  current  in  the  central-office 
relay.  This  current  is  the  net  result  of  current  flowing  in  the  central-office  relay  windings 
from  (1)  the  outlying-point  battery  and  (2)  the  central-office  battery,  due  to  the  duplex 
balance  being  upset  by  the  990-ohm  resistance  in  series  with  the  outlying-point  battery. 

The  variable  resistance  R  at  the  central  office  is  adjusted  so  that  the  spacing  line  battery 
at  the  outlying  point  is  higher  than  the  potential  applied  to  the  apex  of  the  duplex  at  the 
central  office,  thus  insuring  line  current  reversal  when  a  spacing  signal  is  transmitted  from 
the  outlying  point  and  home  copy  if  a  teletypewriter  is  employed. 

Figure  5  shows  a  simplified  type  B  polarential  telegraph  circuit,  in  which  true  polar 
signals  are  transmitted  from  the  central  office,  and  ground  and  +130  volt  battery  are  used 
in  transmitting  marks  and  spaces,  respectively,  from  the  outlying  point.  This  circuit  is 
more  nearly  self-compensating  for  line  leakage  than  the  type  A  circuit. 

When  transmitting  from  the  outlying  point  over  a  dry  (no  leakage)  line,  the  marking 


DIRECT-CURRENT  SYSTEMS 


18-21 


line  current  does  not  affect  the  central-office  receiving  relay,  if  the  balancing  network  at 
the  central  office  exactly  balances  the  line  electrically. 

For  a  spacing  signal  from  the  outlying  point,  aiding  positive  battery,  applied  to  the 
line,  results  in  an  effective  spacing  current  28 /(Ri  -f  R£,  where  E  is  the  outlying  battery 
potential  and  RI  -f-  JtJ2  =  RL  are  as  shown  in  Fig.  5.  The  marking  biasing  current  in 
the  central-office  receiving  relay  is  adjusted  to  a  value  equal  to  E/2Rz,. 


Central  office 


FIG.  5.     Type  B  Polarential  Telegraph  Circuit  (Courtesy  Bell  System) 


During  transmission  over  a  wet  line  with  leakage  Rg  (shown  at  P  on  the  line),  Ra  (at 
the  central  office)  may  be  adjusted  to  maintain  the  potential  at  the  apex  of  the  relay  wind- 
ings at  the  same  value  as  for  the  dry  condition  when  transmitting  a  spacing  signal  from 
the  outlying  point.  Thus,  as  a  result  of  the  compensating  effects  of  the  Ra  adjustment, 
the  received  signals  at  the  central  office  are  not  affected.  If  RI  is  greater  than  2R*,  com- 
plete leakage  compensation  is  not  possible,  unless  the  central-office  battery  voltages  are 
made  higher  than  the  outlying  battery  voltage  or  other  compensation  is  provided.  For 
this  purpose,  Rb  is  provided  for  maintaining  RI  less  than  2R%. 

The  bridge  arm,  with  a  resistance  in  series  with  condenser  C,  is  provided  at  the  outlying 
point  to  neutralize  reverse  current  surges  through  the  receiving  relay  winding  when  the 
outlying  sending  relay  armature  moves  from  space  to  mark.  Such  surges  would  tend  to 
cause  false  breaks  (kick  off)  of  the  receiving  relay  and  mutilation  of  the  home  copy. 

Metallic  telegraph  circuit  operation,  generally  utilizes  telegraph  cable  pairs  or  open 
wires,  or,  when  telephone  facilities  are  involved,  it  is  customary  to  composite  them  to 
secure  the  necessary  telegraph  channels.  The  avoidance  of  interference  between  telegraph 


FIG.  6.    Four-wire  Metallic  Telegraph  Circuit  (Courtesy  Bell  System) 

and  telephone  circuits  and  of  crossfire  between  telegraph  circuits  in  cables  generally  re- 
quires that  the  telegraph  current  values  be  comparable  to  those  on  the  telephone  circuits 
and  that  metallic  circuit  operation  only  be  employed.  Such  operation  may  employ  two 
wires  or  four  wires,  pairs,  side  circuits,  or  phantoms. 

In  four-wire  operation,  as  shown  in  Fig.  6,  separate  paths  are  employed  for  the  two 
directions  of  transmission,  and  artificial  balancing  lines  are  not  required,  as  in  two-wire 


18-22 


TELEGRAPHY 


operation.  These  differences  generally  improve  transmission  over  that  of  the  two-wire 
arrangement,  since  balance  between  the  conductors  and  an  artificial  line  is  not  a  factor. 
Owing  to  the  use  of  four  composited  channels,  as  compared  to  two  such  channels  for  two- 
wire  operation,  the  possibilities  for  conductor  and  equipment  troubles  are  greater  with 
four-wire  operation.  In  general,  four-wire  telegraph  circuits  may  be  superposed  on  two- 
wire  telephone  circuits  of  lengths  ranging  from  about  500  to  1000  miles,  or  on  longer  four- 
wire  telephone  circuits.  Very  long  four-wire  telegraph  circuits  are  not  composited  through- 
out their  entire  length  because  of  prohibitive  low-frequency  delay  distortion  introduced  by 
the  composite  sets. 

Duplex  systems  may  be  of  the  earlier  bridge  type  or  the  later,  more  commonly  used 
differential  type.  Both  these  systems  employ  arrangements  of  telegraph  apparatus  for 
terminating  telegraph  circuits  at  the  central  office  which  permit  the  simultaneous  trans- 
mission of  telegraph  signals  in  both  directions  over  a  single  wire  with  ground  or  metallic 
return  and  without  the  signals  in  one  direction  interfering  with  those  in  the  opposite 
direction. 

Since  bridge  polar  duplex  operation  is  rapidly  being  discontinued  and  is  expected  to  be 
of  little  interest  in  the  future,  its  description  is  omitted  from  this  handbook.  However, 
occasional  reference  is  made  to  it  for  purposes  of  comparison  in  the  following  paragraphs, 
which  relate  to  differential  polar  duplex  operation. 

Figure  7  shows  schematically  the  differential  type  set,  arranged  for  full  duplex  operation. 
The  receiving  relay  has  two  equal  windings,  one  being  connected  in  the  real  line  and  the 

Receiving  relay 

I     Line 


Receiving 
loop 


FIG.  7.    Terminal  Differential  Duplex  Set  Arranged  for  Full  Duplex  Service  (Courtesy  Bell  System) 

other  differentially  in  the  artificial  line  circuit.  Sending  current  from  the  station  battery 
divides  at  the  apex  of  the  receiving  relay  windings  and  flows  equally  in  the  real  and  in  the 
artificial  line  circuits,  assuming  these  are  balanced.  Since  the  receiving  relay  windings 
are  differentially  connected,  the  station  receiving  relay  is  not  affected  by  this  current  flow. 
However,  current  from  a  distant  station  does  operate  the  receiving  relay  at  the  home 
station,  since  it  flows  through  the  windings  of  this  relay  in  a  series  aiding  relation  from 
the  line  through  the  balanced  arms  and  the  artificial  line  to  ground.  Similarly,  the  re- 
ceiving relay  operates  at  the  distant  station  from  incoming  line  current  from  the  home 
station.  Under  these  conditions  full  duplex  operation  is  attained. 

In  the  differential-type  duplex  set,  the  sending  relay  has  both  an  operating  and  a  biasing 
winding.  With  the  sending  loop  key  closed,  the  magnetic  fields  produced  by  these  two 
windings  are  opposing,  but  a  resistance  R  in  series  with  the  biasing  winding  limits  its  cur- 
rent to  about  half  that  in  the  operating  winding  and  the  armature  is  accordingly  held  to 
the  marking  contact  (negative) .  When  the  loop  key  is  opened,  only  the  biasing  winding  is 
effective  and  the  armature  moves  to  its  spacing  contact  (positive) . 

Duplex  systems,  employing  the  ground  for  one  side  of  the  circuit,  are  affected  by  varia- 
tions in  line  constants  due  to  temperature  and  humidity  changes,  but  the  balancing 
artificial  lines  associated  with  each  duplex  set  usually  may  be  adjusted  to  compensate  for 
such  variations.  Duplex  systems  provide  all  the  advantages  of  polar  operation  in  both 
directions  of  transmission,  but  they  also  necessitate  higher-grade  supervision  and  greater 
maintenance  than  some  of  the  simpler  systems,  such  as  in  neutral  operation. 

The  differential  duplex  system  has  largely  superseded  the  older  bridge  duplex  system 
because  of  several  advantages  of  the  differential  over  the  bridge  type.  The  differential 
system  includes  a  simple  differential  duplex  repeater  for  use  at  intermediate  points  on 
long  telegraph  circuits;  in.  the  bridge  system,  two  terminal  sets,  requiring  considerably 


DIRECT-CURKENT  SYSTEMS 


18-23 


more  equipment,  must  be  used  at  such  points.  Intermediate  differential  repeaters  may 
be  employed  on  circuits  having  bridge  polar  type  sets  at  their  terminals,  or  bridge  polar 
intermediate  repeaters  may  be  used-  on  circuits  having  terminal  differential  sets  at  their 
terminals. 

One  of  the  most  important  advantages  of  the  differential  over  the  bridge  system  is  the 
ability  of  its  polar  relays  to  respond  rapidly  to  signals,  particularly  at  the  higher  operating 
speeds.  These  relays,  in  addition 
to  being  more  sensitive  than  the 
relays  ordinarily  used  in  bridge 
polar  sets,  have  a  special  third 
winding  which  forms  part  of  a 
vibrating  circuit  and  materially 
increases  the  relay  response.  The 
vibrating  circuit,  shown,  in  Fig. 
8,  includes  the  third  winding  of 
the  receiving  relay,  whose  arma- 
ture is  connected  to  the  midpoint 
of  the  winding.  One  outer  ter- 
minal of  this  winding  is  grounded 
through  a  condenser,  and  the 
other  outer  terminal  is  grounded 
through  a  resistance.  Batteries 

*«>-«•  m^ota*tnS-WB-wW»^B-i 

contacts. 

Assuming  that  no  current  is  flowing  in  either  the  line  or  artificial  line  windings  of  the 
receiving  relay,  the  relay  armature  moves  back  and  forth  continually  between  the  two 
contacts.  However,  with  current  flowing  in  the  line  and  artificial  line  windings  in  normal 
operation,  the  armature  does  not  vibrate  freely  as  before,  but  the  vibrating  circuit  aids  in 
speeding  up  the  armature  action,  as  the  received  line  signals  cause  it  to  move  between  its 
marking  and  spacing  contacts.  The  free  speed  of  vibration  is  governed  by  the  parameters 
C  and  R,  which  are  usually  adjusted  so  that  the  frequency  of  vibrations  is  slightly  greater 
than  the  line  signal  frequency. 

Half  duplex  operation,  which  provides  for  operation  in  only  one  direction  at  a  time, 
requires  a  means  for  breaking  the  circuit  from  either  terminal  and  for  utilizing  one  loop 
for  both  sending  and  receiving  signals.  Either  the  bridge-  or  differential-type  system  may 
be  employed  for  this  service,  but  each  requires  a  different  circuit  arrangement  of  the  duplex 
set  from  that  required  for  full  duplex  operation.  In  the  bridge-type  system,  the  principal 
additions  required  are  a  control  relay,  holding  coil  for  the  pole  changing  relay,  and  a  re- 
peating sounder;  for  the  differential-type  system  a  control,  break,  and  neutral  relay  are 
added.  The  change  from  full  duplex  to  half  duplex,  or  vice  versa,  for  either  system  is 
readily  accomplished  by  means  of  switches  provided  with  the  duplex  sets. 

Figure  9  shows  a  schematic  diagram  of  a  terminal  differential  duplex  arranged  for  half 
duplex  service.  The  control  relay  functions  on  incoming  signals  to  prevent  the  sending 


2R|         R 

f 

relay      UOJ- 

J 

P 


Loop 


FIG.  9.    Terminal  Differential  Duplex  Set  Arranged  for  Half  Duplex  Service  (Courtesy  Bell  System) 

relay  from  operating  when  the  loop  is  opened  and  closed  by  the  receiving  relay.  Two 
oppositely  poled  batteries  are  connected  to  the  biasing  winding  of  the  sending  and  break 
relays,  and  the  resistances  R  and  2R  are  so  adjusted  that,  with  the  control  relay  contact 


18-24 


TELEGRAPHY 


closed,  the  positive  (spacing)  battery  will  be  in  control  and  the  sending  relay  will  operate 
in  the  same  manner  as  in  full  duplex  operation. 

When  signals  are  being  received  from  the  line,  a  marking  signal  closes  the  loop,  and 
current  flows  from  the  loop  through  the  sending  relay  operating  winding.  When  a  spac- 
ing signal  opens  the  local  loop  at  the  receiving  relay  contacts,  no  current  can  flow  through 
the  sending  relay  operating  winding.  However,  the  control  relay  also  operates  in  unison 
with  the  receiving  relay  (since  their  windings  are  in  series) ,  opening  the  positive  (spacing) 
battery  at  its  armature  contact  and  permitting  negative  (marking)  battery  to  take  con- 
trol of  the  sending  and  break  relays  through  resistance  2R.  Thus,  the  sending  relay  arma- 
ture is  held  on  its  marking  contact  for  either  incoming  marking  or  spacing  signals. 

If  the  local  operator  wishes  to  break  the  circuit  while  incoming  signals  are  being  re- 
ceived, the  local  loop  circuit  is  first  opened  by  opening  the  sending  key.  The  next  spacing 
signal  received  with  the  loop  open  would  be  ineffective,  since  the  signal  would  normally 
open  the  loop  at  the  receiving  relay  contact.  The  next  marking  signal  received  with  the 
loop  open  would  operate  the  receiving  and  control  relays,  and  positive  (spacing)  battery 
would  be  applied  to  the  biasing  windings  of  the  sending  and  break  relays  through  the  con- 
trol relay  armature  contact  and  resistance  R.  Since  there  is  no  current  flowing  through 
the  operating  windings  of  the  sending  and  break  relays  with  the  loop  key  open,  both  relays 
operate  to  spacing.  Positive  battery  is  then  applied  to  the  line  through  the  sending  relay 
spacing  contact,  which  results  in  a  break  signal  to  the  distant  operator. 

The  break  relay  functions  to  insure  a  continuous  break  signal  to  the  line  as  long  as  the 
local  loop  key  is  open,  regardless  of  subsequent  signals  received  from  the  line.  Assuming 
that  one  of  these  signals  is  a  spacing  signal,  the  receiving  and  control  relay  contacts  will 
open,  which  would  permit  negative  (marking)  battery  to  take  control  of  the  sending  and 
break  relays  and  marking  battery  to  be  applied  to  the  line  through  the  sending  relay  mark- 
ing contact  if  a  secondary  circuit  was  not  provided.  Marking  battery  applied  to  the  line 
would  interrupt  the  break  signal.  However,  positive  current  is  maintained  through  the 
biasing  windings  of  the  sending -and  break  relays  as  soon  as  the  break  relay  operates  to 
spacing  (which  it  would  do  on  the  preceding  incoming  marking  signal),  since  positive 
battery  is  then  applied  through  the  neutral  relay,  spacing  contact  of  the  break  relay,  re- 
sistance R,  and  the  biasing  windings  of  the  sending  and  break  relays  to  ground. 

The  neutral  relay  in  this  secondary  circuit  functions  to  prevent  the  possibility  of  the 
circuit  becoming  inoperative  owing  to  the  sending  and  break  relays  at  both  terminals 
becoming  simultaneously  operated  to  spacing.  Under  this  condition,  neither  operator 
could  regain  control  of  the  circuit,  since  the  loop  circuits  at  both  terminals  would  be  open 
at  the  receiving  relay  contacts.  Operation  of  the  neutral  relay  which  occurs  when  the 
control  relay  is  opened  after  the  break  relay  is  operated  to  spacing  short-circuits  the  re- 
ceiving relay  contacts,  so  that  the  sending  and  break  relays  will  be  operated  to  their  mark- 
ing contacts  when  the  associated  loop  key  is  closed. 

The  upset  duplex  method  of  operation  between  a  central  office  and  an  outlying  point 
employs  polar  transmission  generally  from  the  central  office  to  the  outlying  point  and 


Central  office 


Outlying  point 
S         M 


FIG.  10.     The  Upset  Duplex  Method  in  Telegraph  Operation  (Courtesy  Bell  System) 


neutral  transmission  in  the  opposite  direction.  Transmission  in  either  direction  is  affected 
by  line  leakage,  and  transmission  to  the  central  office  is  subject,  in  general,  to  the  same 
limitations  as  for  a  neutral  circuit  employing  a  single  line  repeater.  Because  of  polar 
operation  in  one  direction  and  the  application  of  wave-shaping  units  at  the  outlying  point 
only,  this  method  offers  an  improvement  in  operation  over  the  neutral  circuit  with  a  single 
line  repeater. 

Figure  10  shows  the  upset  method  of  operation,  employing  a  duplex  set,  arranged  for 
half  duplex  operation  at  the  central  office  and  a  teletypewriter  (neutral  operation)  at  the 


DIRECT-CURRENT  SYSTEMS 


18-25 


Schematic  of  Intermediate  Differential  Duplex  Repeater 
(Courtesy  Bell  System) 


outlying  point.  ^The  teletypewriter  normally  operates  on  60  ma  of  line  current  in  an  open 
and  close  loop  circuit,  and  its  line  relay  requires  a  spacing  biasing  current  of  30  ma.  The 
algebraic  sum  of  the  marking  and  spacing  line  currents  while  sending  to  the  outlying 
pouit  must  equal  60  ma,  in  order  to  meet  the  requirements  for  operation  under  the  upset 
method  Also,  when  receiving  at  the  central  office  (neglecting  the  effects  of  series  induct- 
ance and  bridged  capacitance  on  the  line),  the  effective  spacing  current  in  the  receiving 
relay  with  the  outlying  point  key  open  must  equal  the  effective  marking  current  in  this 
relay  with  the  outlying  point  key  closed.  In  practice,  it  is  generally  necessary  to  adjust 
these  spacing  and  marking  currents  in  the  receiving  relay,  since  the  inductance  and 
capacitance  effects  cause  unsymmetrical  wave  shapes  and  cannot  be  neglected. 

The  duplex  repeater  is  employed  at  intermediate  points  in  long  circuits  to  maintain 
proper  operating  currents  in  duplex  telegraph  systems.  The  distance  between  repeater 
points  depends  on  a  number  of  fixed  and  variable  factors,  such  as  the  type  of  line  facilities, 
interference  from  extraneous  sources,  line  leakage,  coordination  of  transmission  levels, 
types  and  quantities  of  central-office  equipment  involved,  line  operating  speeds,  and 
maintenance  considerations. 
Figure  11  shows  schemati- 
cally an  intermediate  differ- 
ential duplex  repeater  circuit 
for  grounded  operation. 

For  60-speed  start-stop 
teletypewriter  operation,  the 
maximum  lengths  of  single 
composited  line  sections  of 
104  mil  and  165  mil  copper 
line  wire,  over  which  it  should 
be  practicable  to  operate 
either  the  differential  duplex 
or  two-path  polar  systems 
without  intermediate  repeat- 
ers, will  be  roughly  within 
the  range  of  170  to  300  miles  and  250  to  450  miles,  respectively.  For  any  given  section, 
the  maximum  length  depends  on  a  number  of  variables,  such  as  the  types  and  quantity 
of  terminal  and  intermediate  equipment  used,  the  line  conditions,  and  maintenance 
schedules  and  type  of  personnel. 

Where  thump,  nutter,  and  crossfire  considerations  permit,  60-speed  differential  duplex 
or  two-path  polar  operation  is  usually  feasible  in  cable  over  a  composited  13  or  16  gage 
wire  with  crossfire  neutralizing  networks  or  over  a  19  gage  simplexed  pair  or  phantom  for 
distances  up  to  about  100  miles. 

The  quadruplex  system  permits  four  messages,  two  in  each  direction,  to  be  transmitted 
simultaneously  over  a  single  grounded  circuit.  One  transmission  in  each  direction  is 
secured  in  the  same  manner  as  has  been  previously  described  under  the  heading  Duplex 
Systems,  article  6.  These  two  channels  and  the  associated  sending  and  receiving  appa- 
ratus are  called  the  polar  side  of  the  quad.  The  two  additional  transmissions  (one  in  each 
direction)  are  obtained  by  varying  the  strength  irrespective  of  the  direction  of  the  line 
current,  and  by  receiving  the  signals,  thus  sent,  on  a  neutral  relay,  which  responds  to 
impulses  of  large  amplitude  irrespective  of  polarity  but  remains  unaffected  by  impulses 
of  smaller  amplitude.  These  two  channels  and  the  associated  equipment  are  designated 
the  neutral  or  common  side  of  the  quad. 

The  schematic  circuit  of  the  differential  quadruplex  is  shown  in  Fig.  12. 

Normally  a  resistance  is  in  series  with  the  pole-changing  relay,  and  a  leak  to  ground  is 
connected  to  the  line,  to  preserve  the  proper  ratio  between  the  marking  and  spacing  cur- 
rent of  the  common  side.  When  this  resistance  is  short-circuited,  and  the  leak  resistance 
removed  by  the  contacts  of  the  transmitting  relay,  in  response  to  the  operation  of  a  send- 
ing key  which  controls  it,  the  line  current  is  increased  to  its  maximum  value.  The  neutral 
relays  in  the  receiving  circuits  are  adjusted  so  that  they  will  respond  to  the  stronger 
signaling  currents  but  will  be  unaffected  by  the  weaker  ones.  Momentary  release  of  the 
neutral  relay  at  the  instant  the  direction  of  line  current  is  reversed  by  operation  of  the  dis- 
tant pole-changing  relay  is  prevented  by  the  discharge  of  a  condenser  through  a  special 
holding  winding. 

The  normal  operating  currents  in  a  quadruplex  set  are  10  to  20  ma  for  the  polar  relays 
and  30  to  60  ma  for  the  neutral  relays.  A  ratio  of  1 :  3  or  1  :  4  between  the  spacing  and 
marking  current  is  required  to  insure  good  operation. 

Although  employed  quite  extensively  at  one  time,  the  quadruplex  system  is  now  used 
but  little,  because  it  was  found  impossible  to  maintain  uninterrupted  operation  of  the 


18-26 


TBLEGEAPHY 


To  Neutral  Side 
Sending  Key. 


FIG.  12.     Differential  Quadruples  Set 


common  side  during  periods  when  damp  weather  or  other  causes  lowered  the  line  insula- 
tion sufficiently  to  reduce  the  ratio  between  the  operating  and  non-operating  currents. 

The  common  side  of  a  quad  is  also  more  susceptible  to  interruption  by  inductive  inter- 
ference from  adjacent  circuits. 

The  bridge  principle,  described  under  duplex  systems,  may  also  be  utilized  in  a  quadru- 
ples system. 

7.  AUTOMATIC  TELEGRAPH  SYSTEMS 

Start-stop  systems,  employing  the  teletypewriter  and  teleprinter  equipment  (similar  type 
units  of  the  Bell  System,  and  Western  Union  Co.,  respectively)  for  sending  and  receiving 
telegraph  signals,  are  so  named  because  of  the  method  of  operation.  This  equipment 
usually  employs  the  seven-element  code,  of  which  the  first  element  is  a  start  pulse,  the 
seventh  element  is  a  stop  pulse,  and  the  other  five  elements  represent  the  character.  Thus, 
the  sending  and  receiving  units  are  synchronized  after  each  group  of  seven  pulses  (ele- 
ments) is  received,  constituting  the  transmission  of  a  character. 

Usually,  the  teletypewriter  or  teleprinter  installation  at  a  subscriber  location  consists 
of  an  electromechanical  unit,  with  or  without  a  keyboard  resembling  that  of  a  standard 
typewriter,  and  having  a  typing  mechanism  for  printing  received  messages  on  a  paper  page 
or  tape.  At  receiving-only  stations,  the  sending  keyboard  is  not  required.  In  some  cases, 
the  subscriber  may  perforate  a  tape,  using  a  keyboard-equipped  perforator,  the  perforations 
representing  coded  characters  of  the  desired  message.  This  tape  may  at  the  same  time  or 
later  be  passed  through  a  tape  transmitter,  which,  being  connected  to  the  line,  sends  pulses 
over  the  line  corresponding  to  the  characters  of  the  message.  Also,  incoming  code  pulses 
may  be  received  by  a  reperforating  machine,  which  perforates  a  tape  with  the  five-element 
code  representing  the  received  characters.  Figure  13  shows  a  sample  of  perforated  tape. 


trerase    -  ?   :  $  S  !  &  JE  8  .,»£ 

Lower  Case    ABQDEFQHi   J  KLMNOPQRSTUVWXYZoSJEJ 


Feed  Ho 

1 

5 

:•        •   •   •                   •    • 
•                   •        «   »    •   •                   • 
.....;..,.%...;.;,,  .;...;.....;. 

:  *•    •:.    •••:•:. 

• 
•t 
•' 

• 
• 

• 
;- 

• 
• 
• 
• 

•  • 

;••• 

• 

• 

• 
• 

• 
•* 

':•' 

FIG.  13.     Teletypewriter  Code  Perforated  in  Tape 

The  teletypewriter  sending  unit  now  in  general  use  consists  principally  of  a  keyboard 
with  key  levers  extending  over  five  notched  selector  bars,  and  a  start-stop  mechanism  of 
driving  and  driven  shaft,  universal  bar,  cams,  eccentrics,  levers,  and  pawls.  Figure  14 
shows  the  general  mechanical  arrangement  of  the  start-stop  mechanism. 

Figure  15  shows  the  details  of  the  key  lever,  which,  when  pressed  down,  positions  the 
selector  bar,  which,  in  turn,  moves  the  locking  latch  head  forward  or  back.  This  latch  will 
either  prevent  the  contact  lever  from  closing  the  transmitter  contacts  (latch  head  forward)  or 
permit  the  contact  lever  to  close  these  contacts '  (latch  head  back) ,  when  the  associated 


AUTOMATIC  TELEGRAPH  SYSTEMS 


18-27 


selector  cam  depression  arrives  at  the  proper  projection  on  the  contact  lever.  Thus,  as  a  key 
lever  is  depressed  to  send  a  particular  character,  the  universal  bar  is  moved  down,  causing 
the  mechanism  (Fig.  14)  to  function  and  the  selector  cams  to  start  rotating.  At  the  same 
time,  the  five  selector  bars  and  locking  latches  are  positioned  by  the  key  lever,  and  each 


Cam  sleeve  assemb 


_,      L       ,       Throw  cut  cam 
Clutch  spring  ,  Drlyen  jay/ 

Driving  ]a 


lutch  lever  eccentric 
'lutch  lever 
Clutch  lever  pawl 


Universal  bar  Selector  bar 

FIG.  14.     Start-stop  Mechanism  of  Teletypewriter  Sending  Unit  (Courtesy  Bell  System) 

of  the  five  contact  levers  is  either  locked  or  left  unlocked,  so  that,  as  the  selector  cams 
make  one  revolution,  each  set  of  transmitter  contacts  does  not  close  or  does  close.  If  the 
contacts  do  not  close,  a  spacing  signal  (no  current)  is  sent  over  the  line,  and  if  the  con- 
tacts close  a  marking  signal  (current)  is  sent  over  the  line. 

Release  of  the  key  lever  after  it  has  been  fully  depressed  causes  the  driven  jaw  to  be 
thrown  out  of  engagement  with  the  driving  jaw  upon  completion  of  one  revolution  of  the 


'lector  cam 


Key  lever 


Selector  bar 


latch 


Key 


Contact  levei 
Transmitter  contai 


Selector 


i lector  cam 


ing  latch 


FIG  15     Positioning  of  Transmitter  Contacts  by  Operating  Key  Lever  of  Teletypewriter  Sending  Unit 

(Courtesy  Bell  System) 

cams,  and  the  rotating  movement  stops-  The  mechanism  is  now  ready  to  send  the  next 
character. 

The  teletypewriter  receiving  mechanism  now  generally  employs  a  single  selector  magnet 
and  a  group  of  six  rotating  cams,  so  spaced  angularly  on  a  shaft  that  each  cam  functions 
at  the  instant  the  corresponding  signal  pulse  is  being  received. 

Figure  16  shows  the  mechanical  arrangement  for  translating  the  selector  magnet  opera- 
tions into  the  positioning  of  the  code  bars.  When  the  open  pulse  is  received,  the  magnet 
armature  releases.  This  operates  a  latch  (not  shown),  allowing  the  cam  shaft  to  rotate. 
The  spacing  of  the  cams  on  the  shaft  is  such  that,  as  the  first  of  the  five  pulses  of  the  code 


18-28 


TELEGRAPHY 


signal  is  being  received,  the  first  cam  engages  the  projection  on  t'he  code  bar  operating 
lever  associated  with  the  first  code  bar  and  rotates  it  slightly  in  a  counterclockwise  direc- 
tion. Assuming  that  the  received  No. 
1  pulse  is  marking,  the  armature  will 
be  operated,  and  the  movement  of  the 
code  bar  operating  lever  will  lift  the 
sword  and  cause  its  upper  right-hand 
projection  to  strike  the  right-hand  end 
of  the  armature  extension.  The  sword 
will  then  rotate  in  a  clockwise  direc- 
tion on  its  pivot  at  A,  so  that,  when 
the  selector  cam  clears  the  code  bar 
operating  lever  and  allows  the  latter's 
spring  to  return  it  to  normal,  the  sword 
point  will  be  forced  against  the  left- 
hand  projection  of  the  T  lever,  rotat- 
ing it  hi  a  counterclockwise  direction 
and  moving  the  No.  1  code  bar  to  the 
right.  If  the  received  No.  1  pulse  had 
been  spacing,  this  code  bar  would  have 
been  moved  to  the  left. 

In  like  manner,  the  other  four  code 
bars  are  properly  positioned  and  the 
sixth  cam  releases  a  clutch,  allowing 
the  printing  mechanism  to  operate,  as 
shown  in  Fig.  17.  The  five  code  bars 
have  been  positioned  to  the  right  or 
left  in  accordance  with  the  five  code 
pulses  received,  which  represent  a  par- 
ticular character.  The  code  bar  slots 

-.      ,-     m  .  .       _      .  .  »,    ,     .          wiU then  *me  UP  so  tnat  the  Putt  bar 

FIG.  16.    Teletypewriter  Receiving  Selector  Mechanism.     .<•      j.u  4.  «u     „„*         -n  «     4.       •     w-i, 
Illustrating  Portioning  of  Code   Bars   (Courtesy  Bell     for  that  character  wiU  center  m  these 
System)  and  only  these  slots  and  will  engage 

the  main  bail.    This  bail  is  then  moved 

forward,  causing  the  pull  bar  to  move  forward  and  the  type  bar  head  to  be  driven  against 
the  paper  or  tape  in°the  teletypewriter. 

After  completion  of  one  revolution  of  the  rotating  cams,  the  rotating  movement  stops, 
owing  to  action  of  the  stopping  mechanism,  and  the  teletypewriter  is  then  ready  to  re- 
ceive the  pulses  for  the  next  character. 


Type  bar 


T  lever 


FIG.  17.   Teletypewriter  Receiving  Mechanism,  Illustrating  Selection  of  Symbol  (Courtesy  Bell  System) 


Power  for  mechanically  operating  both  the  teletypewriter  and  the  teleprinter  is  gen- 
erally secured  from  the  110-volt  commercial  supply  at  their  location,  these  machines 


AUTOMATIC  TELEGRAPH  SYSTEMS 


18-29 


being  equipped  with  either  d-c  motors,  a-c  series  motors,  or  a-c  synchronous  motors.  The 
last  type  of  motor  is  available  for  use  with  50-  or  60-cycle  power  and  is  the  preferable 
type  where  the  power  frequency  has  the  usual  close  regulation.  The  power  consumption 
is  about  115  watts  at  115  volts. 

The  line  signaling  circuit  usually  requires  about  60  to  70  ma  of  direct  current  for  magnet 
and  line  relay  operation,  and  the  remote  control  circuit  (if  provided)  about  50  ma. 

The  regenerative  repeater  may  be  employed  (1)  at  intermediate  points  in  long  com- 
mercial or  private  line  teletypewriter  circuits,  (2)  at  points  where  it  is  desirable  to  divide 
a  teletypewriter  circuit  into  several  sections  because  of  possible  distortion  from  a  large 
number  of  sending  stations  connected  to  the  circuit,  and  (3)  at  teletypewriter  exchanges 
for  switched  connections  and  for  furnishing  service  to  subscriber  teletypewriter  loops 
having  a  relatively  high  transmission  coefficient. 

Start-stop  teletypewriter  signals,  composed  of  one  start  and  five  signal  pulses  of  equal 
unit  length  and  one  stop  pulse  of  1.42  units  length  (teleprinter  signals  are  composed  of 
seven  pulses  of  equal  unit  length),  are  sent  from  the  originating  motor-driven  distributor 
or  other  sending  device  with  mechanical  precision.  However,  as  these  signals  progress 
over  a  telegraph  circuit  several  telegraph  repeater  sections  long,  the  distortion  accumu- 
lates section  by  section,  so  that,  unless  corrected,  the  distorted  signals  received  at  the 
circuit  terminal  would  result  in  message  errors. 

The  regenerative  repeater  located  at  one  or  more  intermediate  points  in  a  long  teletype- 
writer circuit,  as  required,  retimes  and  reshapes  the  received  distorted  signal  pulses  and 
retransmits  them,  as  though  they  were  directly  from  the  originating  machine. 

Figure  18  shows  a  schematic  of  the  circuit  for  one  two-way  regenerative  repeater 
arranged  to  repeat  signals  from  east  to  west  or  west  to  east,  but  not  in  both  directions 


West  receiving  relay       West.east 

regenerator  unit 


Notes: 
@  Indicates  common  130-voIt  120-ohro 

positive  battery  tap. 
Q  Indicates  common  130-voIt  120-ohm    Q 

negative  battery  tap 

FIG.  18.    Schematic  Circuit  of  the  Regenerative  Repeater,  Arranged  for  Half-duplex  Operation  (Cour- 
tesy Bell  System) 

simultaneously.  For  full  duplex  operation,  the  circuit  requires  two  one-way  regenerative 
repeaters.  The  loop  circuits  are  arranged  to  receive  and  transmit  neutral  signals  from 
the  terminal  duplex  telegraph  repeaters  between  which  the  regenerative  repeater  is  usu- 
ally connected.  Each  loop  termination  contains  the  line  winding  of  a  polar  receiving 
relay  and  the  sending  contacts  of  a  regenerator  unit  in  series.  Each  receiving  relay  arma- 
ture controls  a  local  battery  circuit  to  energize  the  magnet  of  the  regenerator  unit  through 
the  marking  contact.  When  the  receiving  relay  armature  moves  to  its  spacing  contact, 
the  receiving  relay  armature  in  the  opposite  loop  is  locked  to  marking. 

The  regenerator  unit  has  both  sending  and  holding  contacts,  the  latter  being  controlled 
by  a  cam  in  such  a  way  that  they  are  open  only  when  the  cam  assembly  is  near  the  stop 
position.  During  retransmission  of  the  signal  elements  of  a  complete  character,  the  hold- 
ing contacts,  being  closed,  apply  battery  of  opposite  polarity  through  a  potentiometer  to 
the  receiving  relay  biasing  winding  of  the  loop,  into  which  the  signals  are  being  retrans- 
mitted. Reversal  of  current  in  this  biasing  winding  holds  its  armature  to  marking,  thus 
preventing  repetition  of  the  retransmitted  signal  back  through  the  regenerative  repeater 
toward  the  originating  terminal. 


18-30 


TELBGEAPHY 


Sendin 


Since  the  length  of  the  retransmitted  signals  is  timed  mechanically  by  the  locking  cam, 
the  outgoing  signal  length  is  normally  independent  of  the  incoming  signal  length.  As 
long  as  that  part  of  the  received  pulse  corresponding  in  time  to  the  release  of  the  magnet 
armature  is  not  affected  by  distortion,  the  outgoing  pulse  will  be  undistorted. 

Tlie  multiplex  system,  a  development  of  1915,  functions  to  divide  the  line  facility  time 
of  a  given  telegraph  channel  (between  two  terminal  points)  among  several  telegraph  cir- 
cuits. The  usual  manually  operated  teleprinter  (or  teletypewriter)  circuit  line  signal  speed 
is  inherently  much  less  than  the  speed  capabilities  of  a  high-grade  telegraph  trunk.  To 
utilize  such  capabilities  efficiently,  the  trunk  time  may  be  allotted  among  two,  three, 
four,  or  more  telegraph  circuits  on  a  full  duplex  basis.  When  four  transmitting  and  four 
receiving  terminals  are  operating  simultaneously  at  66  words  per  minute  over  one  trunk 
line,  this  line  is  handling  528  (8  X  66)  words  per  minute,  thus  enormously  increasing  its 
efficiency  over  single  teleprinter  circuit  operation.  One  multiplex  channel  may  be  oper- 
ated in  each  direction  over  a  duplexed  circuit  or  carrier  channel,  each  of  which  provides 
two  separate  telegraph  paths.  This  system,  because  of  its  high  efficiency  of  operation,  is 
generally  employed  over  heavily  loaded  telegraph  trunk  lines. 

A  code  is  used  in  which  every  character  consists  of  5  equal-length  impulses,  each  of 
which  may  be  either  positive  or  negative,  thus  yielding  32  separate  combinations.  The 
code  as  it  appears  in  a  perforated  transmitting  tape  is  shown  in  Fig.  13.  The  black  dots, 
representing  perforations  in  the  tape,  correspond  to  marking  impulses,  and  the  blank 
spaces  correspond  to  spacing  impulses.  One  of  the  32  possible  combinations,  5  positive 
or  spacing  impulses  in  succession,  has  no  character  assigned  to  it,  and  is  transmitted  con- 
tinuously when  no  messages  are  being  sent.  Two  combinations,  designated  "letters"  and 

"figures,"  are  used  respectively  to  cause  the 
printer  to  print  lower-case  or  upper-case 
characters,  thereby  increasing  the  total  num- 
ber of  characters  that  may  be  transmitted. 

A  schematic  of  this  system  is  shown  in 
Fig.  19.  Each  of  five  contacting  levers  in  a 
transmitter  is  connected  to  correspondingly 
numbered  equal-length  segments  of  a  dis- 
tributor, which  are  successively  connected  to 
the  line  through  the  rotating  brushes  Ff  Fr 
and  an  unsegmented  collector  ring  B.  Dur- 
ing the  time  the  brushes  are  traversing 
another  part  of  the  distributor,  the  trans- 
mitter levers  are  positioned  to  correspond 
with  perforations  in  the  transmitting  tape. 

Receiving) 0,,-^iM  M,M  '*,       After  the  brushes  have  passed  over  those 

segments  and  transmitted  the  signal  com- 
bination to  the  line  the  tape  is  advanced  to 
the  next  set  of  perforations  and  the  cycle 
of  operations  is  repeated.  Auxiliary  or  local  segments  (not  shown)  in  the  distributor  are 
used  to  step  the  transmitter  ahead  at  the  proper  point  in  each  revolution  of  the  distributor. 
The  received  signals  operate  a  polarized  relay,  whose  armature  applies  current  from  a 
local  generator  to  the  brushes  /,  f  of  a  receiving  distributor,  which  comprises  a  solid  col- 
lecting ring  Bf  and  a  segmented  ring  A.' .  Five  segments  of  this  ring  occupying  about  the 
same  angular  position  as  the  sending  segments  are  connected  to  five  correspondingly 
numbered  magnets  which  control  the  operation  of  the  printer.  The  sending  and  receiving 
brushes  at  opposite  ends  of  the  line  are  rotated  in  nearly  exact  synchronism,  and  their 
angular  positions  are  so  adjusted  by  automatic  means  that  the  receiving  brush  will  make 
contact  with  one  segment  at  the  instant  a  pulse  is  received  from  the  similarly  positioned 
sending  segment,  thus  operating  the  corresponding  selecting  magnet.  Therefore,  at  each 
revolution  of  the  brushes  the  five  printer  magnets  will  be  operated  in  sequence  to  repro- 
duce the  identical  combination  set  up  on  the  transmitter  levers. 

One  or  more  additional  sets  of  transmitters  and  printers,  similarly  connected,  provide 
several  independent  channel  transmissions  during  one  complete  revolution  of  the  brushes, 
so  that  by  proper  choice  of  the  number  of  channels  the  full  capacity  of  a  line  may  be  uti- 
lized, while  the  individual  channels  are  operated  at  speeds  that  will  not  exceed  the  capabili- 
ties of  the  operators  or  apparatus. 

The  transmitters  are  controlled  by  a  perforated  sending  tape  (Fig.  13),  prepared  by  the 
operator  on  a  perforator  which  has  a  keyboard  similar  to  that  of  a  typewriter.  Two  types 
of  printers  are  used.  One,  known  as  a  page  printer,  is  similar  to  a  typewriter  and 
is  equipped  with  automatic  means  for  returning  the  carriage  and  for  feeding  the  paper  to 
print  the  message  in  page  form;  the  other  prints  the  message  in  a  continuous  line  on  a 


Distributor  Selecting  Magnets 

FIG.  19.     Schematic  Diagram  of  the  Multiplex 

System 


AUTOMATIC  TELEGRAPH  SYSTEMS 


18-31 


narrow  paper  tape,  which,  is  gummed  on  the  reverse  side  to  permit  of  its  being  readily 
pasted  on  message  forms. 

Distributor  brushes  are  rotated  by  a  synchronous  impulse  motor  of  the  LaCour  type, 
which  is  supplied  with  impulses  of  constant  frequency  generated  by  the  contacts  of  an 
electrically  driven  tuning  fork.  Slight  variations  in  motor  speed  are  compensated  for  by 
applying  a  precise  phase  correction  controlled  by  intelligence  signals  received  from  the 
distant  terminal.  Since  synchronism  is  maintained  between  the  sending  and  receiving 
terminals,  the  standard  start  and  stop  pulses  (of  the  teleprinter)  are  not  required  for  the 
multiplex  circuit,  but  other  local  control  pulses  are  employed. 

Multiples  channels  are  operated  commercially  at  speeds  from  50  to  80  words  per  minute 
(one  word  averages  5  letters  and  a  space,  which  is  equivalent  to  15  cycles  per  word). 
The  number  of  channels  used  is  determined  by  the  traffic  load  and  type  of  circuit  avail- 
able. On  duplexed  land  lines  a  maximum  of  four  channels  in  each  direction  is  permissible. 
Three  channels  can  be  satisfactorily  operated  on  carrier  channels  spaced  170  cycles  apart, 
and  four  multiplex  channels  may  be  applied  on  carrier  channels  spaced  200  cycles  apart. 
Two  to  eight  channels  may  be  successfully  operated  on  submarine  cable,  depending  on 
the  type  of  cable  and  terminal  equipment. 

The  line  frequencies  for  various  speeds  and  number  of  channels  in  each  direction  are 
given  in  Table  1. 

Table  1.    Line  Frequencies 


5-letter 
Words  per 
Minute 

Line  Frequency,  cycles  per  second 

8  Channels 

6  Channels 

4  Channels 

3  Channels 

2  Channels 

50 
60 
70 
80 

100 
120* 
140* 
160* 

75 
90 
105* 
120* 

50 
60 
70 
80 

37.5 
45 
52.5 
60.0 

25 

30 
35 
40 

*  Not  used  commercially. 

Special  repeaters  are  not  required  for  the  operation  of  multiplex  on  duplexed  circuits 
less  than  1000  miles  long,  as  the  regular  duplex  repeaters  are  usually  satisfactory.  On 
longer  circuits,  or  in  those  which  contain  more  than  three  duplex  repeaters  in  succession, 
the  signals  are  likely  to  be  distorted  sufficiently  to  cause  frequent  errors  and  require  exces- 
sive attention  to  the  apparatus.  In  such  circuits  a  regenerative  repeater  is  usually  em- 
ployed at  every  third  or  fourth  repeater  point. 

The  varioplex  system,  a  Western  Union  Telegraph  Co.  development,  is  an  automatic 
telegraph  system  which  provides  for  the  connection  of  up  to  40  individual  telegraph  cir- 
cuits or  subchannels,  having  variable  message  loads,  over  a  single  high-capacity  telegraph 
trunk  circuit,  to  other  individual  telegraph  circuits  or  subchannels  at  one  or  more  distant 
points.  The  actual  number  of  subchannels  operated  over  a  single  varioplex  trunk  is 
limited  mainly  by  practical  considerations,  one  important  factor  being  the  total  message 
load  presented  to  the  trunk  at  any  one  time. 

Though  the  cost  of  this  service  to  the  patron  is  relatively  low,  the  patron  has,  in  effect, 
a  private  high-grade  telegraph  connection  to  the  distant  party.  The  installation  for 
each  patron  consists  of  a  sending  teleprinter  and  a  receiving  teleprinter,  thus  providing 
simultaneous  two-way  service.  Character  counters,  connected  to  the  sending  legs  of 
each  subchannel,  determine  the  number  of  words  sent,  for  billing  purposes. 

Messages  are  transmitted  over  a  given  subchannel  by  operating  the  keyboard  of  the 
sending  teleprinter,  and  messages  are  received  over  a  separate  subchannel  and  automati- 
cally printed  by  the  receiving  teleprinter  on  a  page  or  tape.  Any  patron  may  send  a 
message  at  any  time,  as  desired,  and  the  other  patrons  in  a  given  varioplex  system  may 
do  likewise.  All  the  messages  are  received  at  the  central  office  in  the  varioplex  terminal 
equipment  and  transmitted  in  sequence  over  the  varioplex  trunk  to  its  distant  terminal, 
which  automatically  distributes  each  message  to  the  proper  distant  subchannel,  over 
which  it  reaches  the  patron  for  whom  it  is  intended. 

The  varioplex  terminal  equipment  may  be  classified  as  (1)  individual  to  each  subchan- 
nel, (2)  common  to  all  connections,  and  (3)  part  of  the  varioplex  trunk  circuit. 

Figure  20  shows  a  schematic  of  a  two-channel  (A  and  B)  varioplex  circuit  with^  eight 
patron  offices  and  teleprinters  TPR  at  each  terminal  of  the  circuit  for  one  direction^ of 
transmission  only.  The  opposite  direction  is  similarly  provided.  The  principal  equip- 
ment units  at  sending  terminal  X  for  each  subchannel  include  a  reperforator  RPF,  tape 
transmitter  XTR,  sending  chain  relays  SA  and  &B,  and  sending  control  relay  SCO.  Com- 
mon equipment  units  include  two  banks  (five  relays  per  bank)  of  sending  relays  A  and  B 
and  a  segmented  distributor  sending  ring  (five  segments  per  varioplex  circuit  channel). 


18-32 


TELEGRAPHY 


The  principal  equipment  units  at  receiving  terminal  Y  include,  as  common  equipment,  a 
segmented  distributor  receiving  ring  (matches  the  segmented  sending  ring),  two  banks 
(five  relays  per  bank)  of  receiving  relays  A  and  B,  and  two  local  transmitting  devices  TA 
and  TJB  (each  an  arrangement  of  vacuum  tubes) .  The  individual  subchannel  equipment 
includes  receiving  chain  relays  RA  and  RB  and  receiving  control  relay  RCO. 

Five  sending  and  five  receiving  relays  (banks  A  and  A  or  B  and  B  of  Fig.  20)  are  asso- 
ciated with  the  five  respective  segments  of  each  channel.  These  relays  function  in  such 
a  manner  that,  if  teleprinter  characters  are  being  sent  by  one  or  more  patrons  at  station 
X,  positive  or  negative  potentials  will  be  applied  by  each  relay  to  its  corresponding  seg- 
ment at  station  X,  causing  positive  or  negative  pulses  (combination  determined  by  the 


Varioplex  System 


Station  X 


Station  Y 


FIG.  20.    Schematic  of  Two-channel  (Double-type)  Varioplex  Circuit  (Courtesy  Western  Union  Tele- 
graph Co.,  Electrical  Communication  and  O.  E.  Pierson) 

character  being  sent)  to  be  transmitted  over  the  line  to  station  Y,  as  the  sending  brushes 
pass  over  the  segments  connecting  them,  one  by  one,  to  the  line. 

At  station  Y  the  distributor  brushes  rotate  in  close  synchronism  with  the  distributor 
brushes  at  station  X,  and  each  receiving  segment  is  connected  through  the  line,  for  a 
short  interval  of  time,  to  its  corresponding  sending  segment  on  the  distributor  rings.  As 
the  pulses  are  received  at  station  F,  the  receiving  relays  are  operated  or  released,  depend- 
ing on  whether  the  pulses  are  positive  or  negative. 

The  relay  RA  or  RB  of  a  given  subchannel  will  operate  in  unison  with  the  relay  SA  or 
SB  of  the  corresponding  sending  subchannel  and  will  remain  operated  during  selection  of 
a  character  by  the  receiving  bank  relays  A  or  B.  The  two  relays  RA  and  RB  of  each 
subchannel  control  their  subchannel  circuit  through  their  contacts.  This  circuit  is  closed 
to  a  fixed  potential,  when  both  relays  are  in  their  released  positions,  but  is  disconnected 
from  this  potential  and  is  connected  to  device  TA  or  TB,  upon  operation  of  RA  or  RB, 
respectively.  Devices  TA  and  TB  are  controlled  by  the  contacts  of  the  bank  relays 
A  and  B,  respectively,  and  by  a  segmented  ring  (simplex  ring)  of  the  receiving  distributor. 

Assuming  that  a  given  RA  relay  is  in  its  operated  position,  pulses  from  the  receiving 
simplex  ring  segments  actuate  the  input  circuit  of  the  vacuum  tube  unit  TA  in  accordance 
with  the  selected  character  stored  in  relay  bank  A,  causing  a  signal  combination  to  be 
transmitted  over  the  connected  subchannel  to  the  associated  teleprinter,  which  prints  the 
character  sent. 


AUTOMATIC  TELEGRAPH  SYSTEMS  18-33 

No  distinction  is  made  with  respect  to  which  varioplex  channel  is  used  for  the  transmis- 
sion of  a  character  from  any  given  sending  teleprinter.  Successive  characters  are  sent 
alternately  over  the  two  channels,  one  character  being  supplied  by  each  subchannel  in 
turn.  _  After  all  patrons  who  are  sending  have  transmitted  one  character,  the  cycle  of 
operation  is  repeated.  Successive  characters  from  a  given  subchannel  may  be  trans- 
mitted over  either  varioplex  channel. 

Relays  SA  and  SB  operate  in  a  cyclic  manner,  characteristic  of  the  counting  chain 
type  of  relay  circuit.  A  local  ring  of  the  distributor  furnishes  two  pulses  per  revolution, 
one  pulse  for  operating  each  chain  relay.  Each  chain  relay  is  locked  in  its  operated  posi- 
tion after  being  actuated  by  one  of  these  pulses,  andt  whenever  any  chain  relay  is  actu- 
ated, any  other  previously  operated  relay  in  the  same  vertical  row  (Tig.  20)  is  released. 
Not  more  than  one  SA  or  SB  relay  is  operated  at  one  time.  Also,  the  circuit  is  such  that 
not  more  than  one  relay  can  be  operated  in  the  same  horizontal  row.  If  all  eight  sub- 
channels were  sending  at  the  same  time,  the  chain  relays  would  operate  progressively 
upward,  starting  with  SB  relay  8,  then  SA  relay  7,  SB  relay  6,  and  so  on.  Thus,  in  four 
revolutions  of  the  brush  each  of  the  eight  sending  subchannels  would  transmit  one  char- 
acter. This  cycle  of  operations  would  be  repeated  as  long  as  signals  were  being  sent. 

The  SCO  and  RCO  relays  operate  in  unison,  by  means  of  special  signals  sent  over  the 
line,  to  remove  from  the  operating  chain  circuits  their  associated  SA  and  SB  and  RA  and 
RB  chain  relays,  respectively,  when  these  chain  relays  are  not  functioning.  This  removal 
does  not  affect  the  other  chain  relays,  which  are  operative.  Thus,  by  controlling  the  SCO 
and  R CO  relays,  so  that  they  remain  operative  if  traffic  is  available  at  the  transmitter,  and 
inoperative  if  such  traffic  is  not  offered,  the  main  line  circuit  time  is  made  available  only 
to  those  subchannels  having  a  simultaneous  need  for  it,  and  the  cycle  of  operations  is 
speeded  up.  The  SCO  relay  is  controlled  through  a  collating  arrangement  which  "reads" 
the  character  in  the  transmitter,  causing  the  sending  control  relay  SCO  to  operate  or  re- 
lease, depending  on  the  type  of  character. 

If  the  line  between  stations  X  and  Y  is  capable  of  transmitting  satisfactory  signals  at 
a  rate  of  800  characters  per  minute,  the  two  distributors  for  the  two-channel  varioplex 
circuit  are  adjusted  to  a  speed  of  400  rpm.  This  method  may  be  employed  for  all  high- 
speed automatic  circuits  to  divide  the  total  traffic  capacity  of  a  circuit  into  smaller  and 
more  practical  components. 

Reperforator  switching  systems  are  primarily  switching  arrangements  for  use  in  large 
message  relay  centers.  These  systems  not  only  reduce  to  a  minimum  the  time  required 
in  relaying  messages  through  such  centers  but  also  substantially  increase  the  message 
capacity  of  trunk  lines  and  facilitate  personnel  training  problems.  This  equipment  is 
now  in  operation  in  a  number  of  the  main  telegraph  centers  in  the  United  States,  and 
installations  are  in  progress  in  other  large  centers. 

The  reperforator  switching  system  of  recent  design  consists  principally  of:  (1)  receiving 
equipment,  which  prints  the  incoming  characters  on  and  perforates  them  hi  a  tape;  (2) 
crossoffice  trunkmg  circuits,  by  means  of  which  the  received  message  is  transferred  to  a 
sending  position;  and  (3)  the  sending  equipment,  which  retransmits  the  message  to  its 
destination  or  to  a  second  relay  center. 

Incoming  and  outgoing  transmission  as  handled  by  the  reperforator-switching  system 
is  usually  at  the  rate  of  66  words  per  minute;  the  rate  over  the  crossoffice  trunks  is  150 
words  per  minute,  thus  permitting  rapid  clearance  of  messages  from  the  receiving  posi- 
tions. The  high  rate  over  the  crossoffice  trunk  is  due  to  the  use  of  a  five-wire  trunk,  per- 
mitting the  sending  of  a  five-element  character  at  a  time.  Since  the  attendant  has  only 
two  switch  operations  to  perform  in  order  to  relay  a  message  through  an  office  having 
this  system,  the  time  required  in  handling  the  message  is  only  a  matter  of  seconds.  The 
message  is  passed  through  the  office  entirely  on  tape,  no  manual  receiving  or  sending 
being  involved. 

If  an  outgoing  channel  selected  for  the  transmission  of  a  message  is  idle  at  the  time  of 
selection,  there  is  no  delay  at  this  channel  in  retransmitting  the  message.  If  the  channel 
is  busy  and  no  other  channel  is  selected  for  the  message,  the  tape  from  the  sending  re- 
perforator is  automatically  stored  in  a  specially  constructed  narrow  glass  compartment 
and  is  fed  out  through  the  line  transmitter  automatically,  as  soon  as  the  busy  channel  is 
available. 

Special  centers,  known  as  "spillover"  and  XV  centers,  of  the  switching  office  are  pro- 
vided to  handle  messages  that  are  abnormally  delayed  for  various  reasons,  such  as  circuit 
trouble,  destination  office  closed,  incomplete  address,  and  uncertain  routing,  or  to  handle 
messages  of  an  emergency  or  special  nature.  These  centers  contain  switching  turrets  and 
printer  perforator  receiving  positions  of  the  same  general  type,  as  previously  described. 

Sub  center  switching  systems  (Western  Union  Telegraph  Co.)  are  employed,  where  a 
number  of  branch  offices  and  private  line  patrons  are  localized  in  an  area  some  distance 


18-34  TELEGRAPHY 

from  the  nearest  main  switching  center.  In  order  to  eliminate  delays  in  handling  the 
messages  originating  from  these  sources,  and  to  reduce  operating  costs  which  would  occur 
if  the  messages  were  collected  at  a  local  office  by  messenger  or  otherwise  and  thence  trans- 
ferred  to  a  switching  center,  subcenter  switching  units,  as  required,  are  installed  in  such 
areas  to  extend  the  local  lines  automatically  direct  to  the  main  switching  center  over  a 
small  group  of  trunks.  Usually  the  number  of  trunks  required  is  about  one-third  the 
number  of  local  telegraph  stations  served. 

In  this  sytem,  the  patron's  teleprinter  sends  outgoing  messages  to  the  main  switching 
center  direct,  the  local  line  being  automatically  switched  through  the  subcenter.  A  direct 
circuit  from  the  main  center  to  the  patron  is  established  for  incoming  messages  by  the 
main-center  attendant  dialing  the  patron's  line  number  over  an  idle  subcenter  trunk, 
which  is  automatically  connected  to  the  patron's  line  through  switches  at  the  subcenter. 

In  Teletypewriter  Exchange  Service  (TWX)  in  the  Bell  System,  line  concentrating 
units  are  usually  employed  for  serving  areas,  such  as  described  above.  These  units  are 
arranged  for  automatic  switching  of  two-way  message  service  between  a  group  of  sub- 
scriber lines  at  an  outlying  center  and  a  manual  teletypewriter  switchboard  over  a  small 
group  of  trunks.  One  such  unit  has  a  capacity  of  30  lines,  which  appear  on  the  verticals 
of  three  10  by  10  crossbar  switches.  The  trunks  appear  on  the  horizontals  of  these  switches. 
A  100-line  unit  is  also  available. 

Private  line  switcHng  and  intercommunicating  systems  are  useful  in  extensive  tele- 
graph leased  wire  networks,  such  as  are  employed  to  connect  a  number  of  widely  located 
stations  and  offices  in  large  industries  or  governmental  agencies.  These  systems  perform 
a  service  in  handling  telegraph  messages  somewhat  comparable  to  that  which  private 
branch  exchanges  perform  in  handling  telephone  messages.  The  systems  are  attractive 
to  large  users  because  of  their  simplicity  to  operate  and  freedom  from  trouble. 

The  larger  private  line  switching  systems  of  the  Western  Union  Telegraph  Co.,  receive 
messages  on  printer  perforators,  which  feed  the  perforated  tape  into  a  sending  tape  trans- 
mitter. This  transmitter  is  connected  to  the  desired  outgoing  line  by  means  of  a  plug- 
ended  cord  and  jack. 

When  a  message  is  being  received  at  a  given  unit,  the  attendant  notes  its  destination 
on  the  tape  and  inserts  the  plug  of  the  cord  circuit  associated  with  that  unit  in  one  of 
two  multipled  jacks  in  which  the  outgoing  line  terminates.  Two  jacks  are  provided  for 
each  line  so  that,  if  one  of  the  jacks  is  in  use  and  a  second  message  is  received  for  that  line, 
the  connection  may  be  made  for  this  message  in  the  idle  jack  and  the  message  will  be  sent 
as  soon  as  the  first  message  has  been  cleared. 

Perforator  and  transmitter  units  are  frequently  provided  for  the  patrons,  where  it  is 
desired  to  prepare  messages  in  advance  by  perforating  them  in  a  tape  and  transmitting 
them  at  the  same  time  or  later. 

Fully  automatic  message  switching  systems  in  which  the  first  characters  of  the  code 
serve  to  control  the  switching  equipment  have  been  developed  and  are  now  in  operation. 

Intercommunicating  systems,  as  provided  by  the  Bell  System  for  private  and  govern- 
mental users,  employ  teletypewriter  station  equipment  and  teletypewriter  switchboards, 
at  which  connections  between  stations  or  between  a  station  and  a  trunk  are  established  manu- 
ally by  means  of  cord  circuits.  The  operator  is  provided  with  a  teletypewriter  by  means 
of  which  incoming  calls  are  answered  and  outgoing  connections  are  established  by  typing 
on  a  keyboard.  The  station  may  have  a  page  type  sending-receiving  teletypewriter,  or  a 
sending-receiving  typing  reperforator,  for  printing  characters  on  and  perforating  them  in 
a  tape  (11/i6-in.  wide),  with  which  is  associated  a  transmitter-distributor,  or  other  com- 
binations of  sending  and  receiving  equipment  may  be  provided. 

Teletypewriter  Exchange  service  (TWX),  as  established  throughout  the  United  States 
by  the  Bell  System  for  public  use,  provides  both  large  and  small  switching  centers  inter- 
connected by  trunk  circuits  of  "suitable  grade. 

Thus,  two  TWX  subscribers,  being  provided  with  the  necessary  equipment,  may  com- 
municate with  each  other  by  written  message  from  one  part  of  the  country  to  another 
over  a  vast  network  of  lines  and  equipment,  somewhat  similar  to  that  established  for 
nation-wide  telephone  service. 

Transmission  limitations,  with  respect  to  overall  connections,  are  important  factors  in 
furnishing  a  satisfactory  general  teletypewriter  exchange  service,  as  discussed  in  article  16. 

Teletypesetting,  a  process  of  automatically  setting  type,  is  accomplished  by  perforating 
in  a  tape  the  copy  material  to  be  set  in  type,  and  then  feeding  the  perforated  tape  into  a 
transmitting  device  which  actuates  a  typesetting  machine.  The  tape  may  be  perforated 
locally  or  by  a  perforator  receiving  automatic  signals  of  the  proper  code  over  a  telegraph 
circuit  from  a  distant  point.  The  six-unit  code  is  used  in  the  transmission  of  the  charac- 
ters, and  special  teletypewriter  or  teleprinter  equipment  arranged  ta  send  and  receive  this 
code  is  employed,  providing  page  copy  for  checking  purposes. 


ALTERNATING-CURRENT  TELEGRAPH  SYSTEMS      18-35 


8.  ALTERNATING-CURRENT  TELEGRAPH  SYSTEMS 

Alternating-current  telegraph  systems  employ  both  wire  and  radio  channels  for  the 
transmission  of  telegraph  signals. 

Voice-frequency  carrier  telegraph  systems  operate  within  the  lowest  carrier  frequency 
(voice)  range  of  about  250  to  3150  cycles. 

One  such  system,  now  in  common  use,  provides  up  to  IS  two-way  telegraph,  channels 
having  carrier  frequencies  from  255  to  3145  cycles,  spaced  170  cycles  apart.  This  particu- 
lar system  operates  on  a  four-wire  line  basis  over  various  types  of  facilities,  such  as  loaded 
four-wire  cable  circuits,  open-wire  physical  circuits  on  a  four-wire  basis  (noise  conditions 
permitting),  and  the  different  types  of  carrier  telephone  channels  under  suitable  condi- 
tions. 

Figure  21  shows  the  principal  elements  of  a  telegraph  channel  from  the  sending  to  the 
receiving  telegraph  terminal  for  the  particular  eighteen-channel  system  referred  to  above. 
At  the  sending  terminal,  the  marking  signal  closes  the  line  circuit  at  the  step-up  trans- 
former and  the  carrier  current  is  transmitted  over  the  line  to  the  receiving  terminal,  where 
it  is  amplified  and  detected,  and  finally  operates  the  receiving  relay  to  marking  <  — ) .  For 

Sending  Receiving         Amp.  det.     Receiving 

filter  filter  level  comp.       relay 


D-c  telegraph 
circuit 

-<s— 1 

From  d-c 
telegraph  circuit 

Sending  relay 

PIG.  21.     Principal  Elements  of  a  Carrier  Telegraph  Channel  (Co-urtesy  Bell  System) 

the  spacing  signal,  the  input  of  transformer  T2  is  short-circuited  at  the  step-up  transformer, 
and  no  carrier  current  reaches  the  line.  The  receiving  relay  is  operated  to  spacing  (4), 
when  no  current  is  being  received  from  the  line.  The  0.2-megohm  resistance  inserted  in 
series  with  the  line  winding  of  transformer  TI  assists  in  reducing  the  carrier  to  the  line 
and  lessening  the  load  on  the  carrier  supply  during  spacing  in  this  system.  The  send  and 
receive  filters  pass  only  the  carrier  frequency  assigned  to  the  particular  channel  with 
which  they  are  associated. 

The  telegraph  level  at  a  given  point  on  the  circuit  is  the  power  at  that  point  due  to  a 
single  steadily  marking  channel  and  is  expressed  in  decibels  referred  to  1  milliwatt  (dbm). 
If  frequency  distortion  is  present,  the  level  refers  to  a  channel  operating  at  or  near  1000 
cycles,  which  is  the  nominal  telegraph  level.  The  telegraph  levels  on  the  line  section  of  a 
circuit,  in  general,  must  be  high  enough  to  override  line  interference  and  low  enough  to 
prevent  modulation  and  crowding,  which  introduce  objectionable  interference  in  other 
systems.  The  sending  and  receiving  levels  in  the  same  telegraph  system  should  be  such 
that  cross-induction  will  not  occur  between  the  sending  and  receiving  branches  in  ihe 
terminal  cabling  and  equipment. 

Specific  telegraph  level  (STL)  of  a  circuit  used  alternately  for  telephone  service  is 
numerically  equal  to  the  power  of  one  telegraph  channel  (in  dbm)  at  a  point  of  zero  tele- 
phone transmission  level.  At  present  in  the  Bell  System,  this  level  is  — 16  db  for  type  C 
carrier  (except  in  certain  cases)  and  —  21  db  for  other  telephone  facilities.  The  proper 
selection  of  the  STL  in  any  given  case  is  highly  important  in  the  application  of  telegraph 
to  telephone  circuits,  because  of  the  dependency  of  the  satisfactory  operation  of  both 
services,  when  related,  on  this  factor. 

The  telegraph  signaling  spaed  that  can  be  obtained  on  a  carrier  channel  depends  on  the 
band  width  or  frequency  range  of  the  channel.  With  wider  bands,  higher  signaling  speeds 
per  channel  are  possible  but  fewer  channels  are  obtained.  When  the  carrier  channels  are 
spaced  170  cycles  apart,  each  channel  will  allow  telegraph  signaling  at  speeds  of  35  to  40 
cycles  per  second,  which  permits  of  the  operation  of  a  three-channel  multiplex  working  at 
50  words  per  minute  per  multiplex  channel.  Thus,  a  twelve-channel  carrier  system  oper- 
ated in  this  way  would  have  a  total  capacity  of  1800  words  per  minute  in  each  direction 
simultaneously.  With  300-cycle  spacing  between  carrier  channels,  telegraph  speeds  as 


18-36  TELEGRAPHY 

high  as  75  cycles  per  second  may  be  obtained  on  each  channel,  which  permits  the  operation 
of  a  four-channel  multiplex  at  75  words  per  minute  per  multiplex  channel.  An  eight- 
channel  system  of  this  type  has  a  capacity  of  2400  words  per  minute  in  each  direction 
simultaneously. 

Voice-frequency  carrier  systems  are  usually  operated  over  four-wire  circuits,  one  pair 
being  used  for  the  channels  working  in  one  direction  and  the  remaining  pair  for  channels 
working  in  the  opposite  direction.  They  may  be  operated  on  two-wire  circuits,  however, 
by  transmitting  the  lower  half  of  the  carrier  frequencies  in  one  direction  and  the  upper 
half  in  the  opposite  direction  in  a  manner  similar  to  the  high-frequency  carrier,  which  is 
described  in  this  article. 

Standard  voice  or  carrier  telephone  repeaters  are  employed  for  voice-frequency  carrier 
telegraph  systems  assigned  to  voice  or  carrier  telephone  facilities.  These  repeaters  take 
the  same  spacing  as  would  be  employed  if  the  facilities  were  assigned  to  telephone  circuits. 

Carrier  supply  for  the  different  carrier  telegraph  frequencies  in  the  type  of  system 
shown  in  Fig.  21  is  now  generally  furnished  by  vacuum-tube  oscillators,  although  some 
motor-driven  multifrequency  generator  sets  are  still  in  service.  One  vacuum-tube  oscil- 
lator unit  has  a  capacity  for  as  many  as  50  carrier  telegraph  channels  normally  but  for  a 
short  time  may  supply  a  much  greater  number  of  channels.  Harmonic  control  is  provided 
for  the  vacuum-tube  oscillator  supply  to  limit  high  peak  line  currents,  which  would  tend 
to  overload  the  telephone  repeaters  and,  in  carrier  systems,  cause  interchannel  inter- 
ference. 

Voice-frequency  carrier  transmission  is  sometimes  employed  between  a  teletypewriter 
office  and  an  outlying  teletypewriter  station,  particularly  where  a  d-c  telegraph  channel 
cannot  be  satisfactorily  provided. 

In  this  system,  a  carrier  frequency  of  690  cycles  transmitting  from  the  office  and  1640 
cycles  transmitting  from  the  station  is  used.  Carrier  current  is  transmitted  for  spacing 
and  no  current  is  transmitted  for  marking  signals.  The  break  relay  at  the  office  can  thus 
be  omitted,  since  the  sending  relay  opens  the  receiving  circuit  during  transmission  of  a 
spacing  signal,  which  holds  the  receiving  relay  on  its  marking  contact.  Also  carrier 
current  is  not  sent  over  the  line  from  either  terminal  while  the  receiving  circuit  at  that 
terminal  is  connected  to  the  line. 

Filters  are  required  at  both  terminals,  in  order  to  prevent  echo  current  effects  from 
reaching  the  detectors  and  causing  false  signals. 

The  series  resistance  across  the  low-pass  filter  at  the  station  provides  for  spacing  signal 
feedback  to  the  station  amplifier  detector  circuit  and  permits  obtaining  a  home  copy  of 
the  outgoing-station  message. 

The  operation  of  this  carrier  system  does  not  affect  the  usual  telephone  circuit  trans- 
mission over  the  line  and  is  not  materially  affected  by  reasonable  amounts  of  line  leakage 
or  by  earth  potentials,  crossfire,  or  power  induction  in  the  circuit. 

High-frequency  carrier  telegraph  systems  for  open-wire  application  have  been  employed 
by  the  larger  communication  companies  for  a  number  of  years.  The  ten-channel  open- 
wire  systems  of  the  Bell  companies  are  no  longer  standard,  owing  to  the  improvements 
and  economies  secured  from  the  eighteen-channel  voice-frequency  system  described  above. 

The  Western  Union  Telegraph  Co.  has  developed  a  series  of  carrier  telegraph  systems 
specially  designed  to  fill  the  requirements  of  the  domestic  telegraph  system.  A  number 
of  these  systems  are  in  operation  (1947),  and  additional  systems  are  being  installed  rapidly 
in  a  modernization  program  in  which  carrier  operation  will  replace  d-c  telegraph  opera- 
tion for  trunk  circuits  and  will  substantially  reduce  the  number  of  wires  and  pole  lines 
required  for  the  telegraph  service.  These  systems  include  (1)  a  portable  type  for  estab- 
lishing short  temporary  or  emergency  channels,  (2)  a  low-cost  four-channel  system  for 
distances  not  greater  than  two  repeater  sections,  and  (3)  four  types  of  multichannel, 
long-distance  trunk  systems.  The  basic,  standard  unit  of  the  trunk  carrier  systems  is  a 
3000-cycle  voice-frequency  band  extending  from  300  to  3300  cycles.  The  four  trunk 
systems  are  (1)  the  7.5-kc  type  E  with  one  voice-frequency  band  in  each  direction,  (2)  the 
15-kc  type  F  with  two  bands  in  each  direction,  (3)  the  30-kc  type  G  with  four  bands  in 
each  direction,  and  (4)  the  type  WN  with  32  bands  in  each  direction  and  requiring  a 
transmission  band  of  150  kc  in  each  direction.  Types  E,  F,  and  G  are  open-wire  systems 
designed  for  two-wire  operation  and  with  frequency  allocations  (for  300-cycle  channels) 
as  shown  in  Fig.  22.  The  type  WN  is  designed  for  use  with  a  two-way  microwave  radio 
relay  circuit. 

In^multiband  systems,  only  one  band  can  be  transmitted  in  its  original  frequancy 
position  (300  to  3300  cycles) ;  all  other  bands  must  be  transferred  or  translated  to  separate 
positions  hi  the  available  frequency  spectrum  of  the  transmission  medium.  This  can  be 
accomplished  by  modulating  each  of  the  bands  requiring  translation  by  a  separate  second- 
ary carrier  of  appropriate  frequency.  Such  a  method  is  wasteful  of  the  frequency  spec- 


ALTEKNATING-CTJRRENT  TELEGRAPH  SYSTEMS      18-37 


trum  because  of  the  increasing  inefficiency  (in  absolute  band  width)  of  selective  filters  as 
the  frequency  is  increased.  The  Western  Union  systems  utilize  the  spectrum  more  effi- 
ciently by  employing  a  plural  or  tandem  method  of  modulation  in  which  the  voice-fre- 
,  quency  bands  are  translated  in  groups,  thus  requiring  only  broad  group  niters  at  the 
assigned  position  in  the  frequency  spectrum.  The  secondary  carrier  or  translation  fre- 
quencies are  supplied  by  amplified  harmonics  of  a  high-stability,  low-frequency  oscillator. 

Although  these  carrier  systems  are  designed  primarily  for  operation  over  Western  Union 
facilities,  they  are  readily  adapted  to  operation  on  any  wire  or  radio  transmission  system 
that  provides  a  suitable  transmission  band.  Furthermore,  the  individual  voice-frequency 
bands  can  be  repeated  or  patched  at  will  between  the  four  types  of  Western  Union  multi- 
band  systems,  Bell  System  facilities,  and  equivalent  facilities  of  other  companies. 

The  voice-frequency  bands  may  be  utilized  for  high-speed  facsimile  telegraph  trans- 
mission or  for  operation  as  a  telephone  circuit,  but  they  are  ordinarily  channelised  for 


E  SYSTEM 


F  SYSTEM 


68  0246 

Frequency  in  kilocycles 

Q   SYSTEM 


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6.86  K.C.  carrier        114.06  K.C-.  carrier 

and 


23.66  K.C.  pilot       80.86  K-C.  carrier 
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pilot  channel 

„ ocations  for  ' „.._., . 

Union  Telegraph  Co.,  A.I.E.E.,  F.  E.  D'Humy,  and  P.  J.  Howe) 


FIG.  22.     Frequency  Allocations  for  Three  Types  of  ^Carrier  Telegraph  Systems  ^(Courtesy  Western 

""  '  .E.E.,      F.      "          T^'  T      T-W  T          TT 


printer  or  other  code  telegraph  operation.  Two  types  of  channelization  are  employed, 
providing  (1)  nine-wide-band  channels  spaced  300  cycles  apart  and  suitable  for  four- 
channel  multiplex  operation  at  speeds  up  to  75  cycles  per  second,  and  (2)  eighteen,  narrow- 
band channels  spaced  150  cycles  apart  and  suitable  for  teleprinter  or  two-channel  multi- 
plex operation.  Present  trends  are  away  from  the  relative  mechanical  complexity  of 
multiplex  telegraphy  and  toward  teleprinter  operation;  consequently,  narrow-band 
channels  only  are  now  being  installed. 

The  following  description  deals  with  narrow-band  channels,  but  it  also  applies  gen- 
erally to  the  basically  similar  wide-band  equipment.  The  sending  terminal  of  each  chan- 
nel includes  a  vacuum-tube  oscillator,  which  supplies  the  channel  carrier  frequency.  The 
d-c  telegraph  signals  are  impressed  or  superposed  on  the  carrier  by  modulation.  The 
carrier  telegraph  systems  of  other  companies  use  amplitude  modulation,  as  did  the  early 
Western  Union  systems.  For  amplitude-modulated  (a-m)  operation,  the  channel  carrier 
frequency  is  transmitted  for  a  marking  signal  and  interrupted  for  a  spacing  signal.  West- 
ern Union  has  introduced,  and  now  uses  exclusively,  frequency  modulation  for  carrier 
telegraph  transmission.  In  f-m  transmission,  the  frequency  of  the  channel  oscillator  is 
controlled  by  the  d-c  telegraph  signals  being  transmitted.  Marking  and  spacing  signals 
cause  the  frequency  to  swing  down  and  up,  respectively,  35  cycles  from  midchannel  fre- 
quency. The  transitions  between  marking  and  spacing  frequencies  are  not  instantaneous 
but  are  very  abrupt,  with  the  frequency  increasing  and  decreasing  smoothly  between  the 
lower  and  upper  limits.  This  action  results  in  true  frequency  modulation,  and  the  modu- 
lated carrier  current  appears  at  the  output  of  the  channel  transmitting  filter  constant  in 
amplitude  but  varying  in  frequency. 

At  the  distant  corresponding  channel  terminal,  the  carrier  is  selected  by  the  receiving 
filter  and  amplified,  after  which  it  passes  through  an  amplifier  limiter,  a  frequency  dis- 
criminator, and  a  rectifier  circuit,  which  restores  the  marking  and  spacing  frequencies  to 


18-38  TELEGRAPHY 

their  original  d-c  form.  The  vacuum-tube  limiter  circuit  maintains  a  constant-amplitude 
output  for  received  levels  in  a  range  between  —50  and  +10  dbm. 

In  comparison  with  a-m  transmission,  f-m  transmission  is  definitely  less  susceptible  to 
interference,  and  the  received  signals  are  immune  to  the  influence  of  level  variations  en- 
countered in  transmission. 

The  original  f-m  channel  terminals  contained  transmitting  and  receiving  relays.  In 
the  latest  type  of  narrow-band  equipment,  the  relays  have  been  eliminated  and  operation 
of  the  channel  terminals  is  completely  electronic.  This  change,  together  with  other 
simplifications,  has  effected  a  considerable  reduction  in  size  and  cost  of  equipment. 

Transmission  losses  in  the  open-wire  systems  between  sending  and  receiving  terminals 
are  compensated  for  by  vacuum-tube  repeaters,  spaced  at  suitable  intervals  along  the 
line.  Average  wet-weather  losses  are  limited  to  about  25  db  between  repeaters.  Reflec- 
tion losses  at  junctions  of  open  wire  and  incidental  intermediate  or  terminal  cables  are 
limited  by  impedance-matching  networks.  Loading  of  short  cables  to  reduce  attenuation 
losses  is  no  longer  favored.  Networks  are  provided  at  repeaters  and  terminals  to  equalize 
the  losses  at  different  frequencies,  so  that  the  attenuation  is  substantially  uniform  over 
the  carrier  band. 

Telegraph  circuits  can  be  operated  across  the  United  States  entirely  by  carrier  without 
requiring  a  regenerative  repeater  at  any  point.  The  type  G  system  provides,  in  each 
direction,  on  a  single  pair  of  wires  72  teleprinter  circuits  or  144  multiplex  channels  having 
a  total  capacity  of  4750  and  9500  words  per  minute,  respectively,  in  each  direction.  The 
other  systems  provide  capacities  proportionate  to  the  number  of  voice-frequency  bands 
employed.  The  radio  relay  system  with  its  576  carrier  channels  handles  38,000  words 
per  minute  each  way. 

9.  FACSIMILE  SYSTEM 

Facsimile,  as  defined  and  discussed  in  detail  elsewhere  in  this  handbook,  is  a  process 
whereby  such  objects  as  a  picture  or  a  sheet  of  paper  containing  printing  or  writing  are 
electrically  scanned,  and  the  electrical  currents  thus  generated  are  transmitted  by  wire  or 
radio  to  a  receiving  device  which  reproduces,  as  a  print  or  on  a  specially  prepared  paper, 
the  original  picture  (in  black  and  white)  or  the  printing  or  writing.  The  following  brief 
discussion  applies  to  facsimile  as  used  in  telegraph  operation  only. 

Facsimile  transmission  of  messages,  as  employed  by  the  Western  Union  Telegraph  Co., 
is  known  as  telefax  service  or  transmission.  This  type  of  service  was  considered  desirable 
by  the  telegraph  companies  for  many  years,  but  the  recording  systems  used,  employing 
photographic,  chemical,  or  other  processes,  were  too  costly  or  slow  for  the  general  handling 
of  telegrams. 

In  recent  years,  the  Western  Union  Telegraph  Co.  developed  for  its  use  a  facsimile 
recording  paper,  with  the  trademark  Teledeltos.  This  paper  has  a  conducting  coating  of 
a  light  gray  color  which  is  marked  in  black  by  the  passage  of  an  electric  current  through 
it.  The  paper  is  used  dry,  requires  no  processing,  is  not  affected  by  light  or  ordinary 
moisture  conditions,  and  produces  immediately  a  clear-cut  permanent  record.  Simple 
Telefax  recording  equipment  receives  the  incoming  amplified  facsimile  currents  and 
applies  them  directly  to  the  paper  by  means  of  a  stylus  riding  on  the  surface  of  the  paper. 

The  Telefax  equipment  thus  far  developed  is  of  several  different  types  and  has  been 
designed  primarily  to  provide  maximum  convenience  for  the  general  public.  However, 
trunk-line  Telefax  is  being  used  to  some  extent  between  large  centers,  as  between  Chicago 
and  New  York.  These  trunk  circuits  have  been  used  mainly  for  the  transmission  of 
drawings,  sketches,  copy  for  publication,  editorial  corrections,  and  commercial  messages. 
For  good-quality  Telefax  transmission,  a  frequency  band  width  of  about  2500  cycles  is 
required.  The  line  loss  at  the  maximum  frequency  used  should  not  exceed  about  25  db. 

10.  MISCELLANEOUS  TRANSMITTING  AND  SIGNALING  SYSTEMS 

The  ticker  system  is  designed  to  furnish  stock,  bond,  and  grain  quotations  and  pertinent 
news  items  to  brokerage,  investment,  and  private  offices  during  trading  hours. 

The  ticker,  which  does  not  include  a  transmitting  keyboard,  is  otherwise  quite  similar 
in  principle  to  the  start-stop  printer,  the  chief  points  of  difference  being  in  the  form  of 
type  wheel  used  and  the  method  employed  in  shifting  from  letters  to  figures.  An  eight- 
unit  code  is  used.  The  first  impulse  of  each  group  starts  the  printer;  the  second  impulse 
determines  by  its  polarity  whether  letters  or  figures  are  to  be  printed;  the  succeeding  five 
pulses  determine  the  particular  character  to  be  printed;  and  the  eighth  pulse  is  the  stop 
pulse,  which  allows  the  distributing  mechanism  to  come  to  rest  in  preparation  for  the 


MISCELLANEOUS  SIGNALING  SYSTEMS 


18-39 


next  succeeding  signal  group.  The  tickers  are  controlled  by  a  single  transmitter  at  the 
central  distributing  point.  Normal  operating  speed  is  from  450  to  500  printed  characters 
per  minute. 

The  teleautograph  system  is  employed  principally  by  banks,  railroads,  department 
stores,  and  similar  businesses.  This  system  provides  a  means  by  which  messages,  written 
in  longhand  at  one  station,  may  be  reproduced  simultaneously  at  one  or  more  stations  at 
various  locations  without  material  distortion  of  the  original  characters. 
^  The  principle  on  which  this  system  operates  is  shown  in  Fig.  23.  The  transmitter  con- 
sists of  a  stylus,  which  is  mechanically  connected,  through  two  sets  of  levers  and  appro- 
priate swivel  joints,  to  the  contact  arms  of  two  variable  rheostats  in  such  a  way  that  the 
horizontal  and  vertical  components  of  the  stylus  movements  are  translated  into  cor- 
responding current  variations  in  two  lines  connecting  the  receiver.  At  the  receiver,  the 
variations  in  the  line  currents  produce  similar  movements  in  two  coils  or  "buckets" 
within  a  magnetic  field.  The  movements  of  these  coils  are  communicated  through  a 
system  of  levers  to  a  writing  pen  which  reproduces  the  movements  of  the  sending  stylus. 


[  Relay"  Paper  Shifter      Vlb.  (p  L)  Reh 

L     1         S.gn.lBuzz.r/  /      ^.^ 


Transmitter 
FIG.  23.     Schematic  Diagram  of  the  Telautograph  System 

The  pen  is  lifted  from  the  paper  and  the  paper  is  shifted  to  provide  fresh  writing  surface 
by  magnets,  which  are  controlled  by  superposed  alternating  current  supplied  by  a  buzzer 
located  in  the  transmitter. 

Two  grounded  circuits  are  required,  one  to  transmit  the  current  variations  representing 
each  of  the  two  components  of  the  stylus  motion.  As  many  as  100  receivers,  arranged  to 
be  controlled  by  one  or  more  transmitters,  may  be  operated  in  multiple  on  a  pair  of  wires. 
The  speed  of  operation  is  determined  entirely  by  the  rapidity  with  which  the  operator 
can  write.  The  line  potential  is  usually  120  volts  with  one  side  grounded,  and  current 
depends  upon  the  resistances  of  transmitters  and  receivers.  One  type  of  equipment, 
which  normally  operates  on  100  ma,  is  adapted  for  use  on  wires  not  exceeding  700  ohms 
resistance  each.  Another  type  requires  only  60-ma  line  current  and  will  operate  well  on 
lines  up  to  1200  ohms  resistance,  although  in  a  few  extreme  cases  operation  has  been 
maintained  on  lines  having  resistances  as  high  as  1800  ohms. 

Messenger  call  circuits  are  employed  for  summoning  telegraph  messengers  to  pick  up 
messages  at  various  locations  and  deliver  them  to  the  telegraph  office  for  sending. 

Signals  are  transmitted  by  a  call  box,  which  consists  of  a  spring-driven  clockwork 
mechanism  arranged  to  turn  a  pair  of  notched  contact  wheels  through  one  complete  revo- 
lution each  time  the  box  is  operated  by  turning  and  releasing  a  winding  key,  as  shown  in 
Fig.  24.  The  contact  wheels  are  arranged  both  to  open  and  ground  the  line  when  trans- 
mitting signals,  and  to  restore  the  line  to  its  normal  closed  ungrounded  condition  upon 
coming  to  rest.  The  call  boxes  are  connected  in  series  with  a  line  having  both  ends  ter- 
minated at  the  central  office  in  relays  which  operate  a  buzzer  and  a  register  for  recording 
the  signals  on  a  paper  tape.  Switches  Si,  &,  and  S$  are  provided  to  change  the  line, 
battery,  and  register  connections  to  permit  of  receiving  signals  from  the  call  boxes  even 
during  times  when  the  line  circuit  may  be  accidentally  open  or  grounded  or  both.  Only 
the  simultaneous  occurrence  of  a  fault  on  both  sides  of  a  call  box  or  group  of  boxes  will 
prevent  the  transmission  and  correct  reception  of  signals.  The  normal  line  current  in 
call  circuits  is  50  ma,  and,  for  satisfactory  service,  not  more  than  50  call  boxes  should  be 
included  in  any  one  circuit.  Signaling  speed  usually  does  not  exceed  4  impulses  or  2 
cycles  per  second. 


18-40 


TELEGRAPHY 


Though  many  technical  advances  have  been  made  in  the  art  of  telegraphy,  the  mes- 
senger call  box,  of  which  there  are  over  300,000  in  the  United  States  in  offices,  hotels,  and 
other  public  places,  still  plays  an  important  part  in  the  collection  of  commercial  telegraph 
messages. 


FIG.  24.    Schematic  Diagram  of  a  Messenger  Call  Circuit 

Clock  circuits  are  employed  in  furnishing  telegraph  time  service.  The  clocks  at  sub- 
scriber premises  are  driven  by  springs  which  are  wound  periodically,  usually  once  each 
hour,  by  a  small  electric  motor  operating  on  dry  batteries.  The  minute  and  second  hands 
are  arranged  so  that  they  may  be  moved  to  their  12  and  60  positions,  respectively,  by  the 
operation  of  a  synchronizing  magnet.  The  synchronizing  magnets  of  the  clocks  are  con- 
nected in  series  in  a  grounded  circuit  terminating  at  the  central  office  on  the  contacts  of  a 
transmitting  relay  or  a  synchronizing  machine  which  sends  a  synchronizing  impulse  1  sec 
long  once  every  hour,  in  response  to  the  operation  of  contacts  of  a  master  dock.  The 
transmitting  circuit  is  arranged  to  give  an  audible  signal  if  the  synchronizing  impulse  fails 
to  be  transmitted  owing  to  line  failure.  The  clocks  mechanically  lock  their  synchronizing 
mechanisms  in  the  inoperative  position  except  for  two  or  three  minutes  immediately 
preceding  and  following  the  time  at  which  the  synchronizing  impulse  is  to  be  received,  so 
as  to  protect  the  clocks  from  being  set  to  a  false  position  by  accidental  crosses  between 
the  line  and  power  circuits.  The  normal  line  current  is  250  ma,  and  approximately  60 
clocks  may  be  operated  on  one  circuit. 

Naval  Observatory  time  signals  are  regularly  distributed  to  all  parts  of  the  United 
States  by  the  Western  Union  Telegraph  Co.  over  about  200,000  miles  of  wire  network. 
These  signals  provide  the  means  for  maintaining  some  2000  master  clocks,  so  that  they 
continuously  indicate  time  to  a  practical  degree  of  accuracy. 

Railroad  communication  systems  employ  the  latest  types  of  telegraph  as  well  as  tele- 
phone and  radio  facilities  for  the  control  of  train  movements  and  the  general  business  of 
the  railroads.  The  telegraph  facilities  used  are,  in  general,  similar  to  those  previously  dis- 
cussed in  this  section  and  will  not  be  considered  further  here.  Special  arrangements  of 
telegraph  facilities  are  employed  by  the  railroads  to  meet  their  special  needs. 


SUBMARINE  CABLE  TELEGRAPHY 

By  John  D.  Taylor 


Submarine  cables  interconnect  all  the  earth's  continents  for  the  transmission  of  tele- 
graph messages. 

One  large  telegraph  company  has  more  than  30,000  miles  of  ocean  cable,  some  of  it  lying 
at  a  depth  of  nearly  3  miles.  This  company  laid  its  first  cable  in  1873  and  its  latest  cable 
in  1928.  The  North  Atlantic  is  spanned  by  14  cables,  and  6  cables  extend  between  North 
America  and  the  Azores,  where  connections  are  made  with  Europe  and,  via  the  Cape 
Verde  Islands,  with  Africa  and  South  America.  There  are  other  cables  between  the 
United  States  and  Mexico,  South  America,  and  the  West  Indies.  Three  cables  span  the 
Pacific  Ocean. 


CABLE  DATA 


18-41 


11.  CABLE  DATA 

The  first  transoceanic  cables  laid  were  of  the  non-loaded  type.  It  was  not  until  1924 
that  the  first  loaded  cable  was  placed,  and  this  cable  connected  New  York  with  Horta, 
Azores  Islands. 

Non-loaded  cables  of  the  deep-sea  type  usually  consist  of  a  single  copper  conductor, 
which  may  be  solid,  stranded,  or  a  combination  of  both,  to  provide  greater  flexibility,  as 
shown  in  Fig.  1.  This  conduc- 
tor is  encased  in  gutta-percha  * 
insulation,  covered  with  servings 
of  jute  yarn  over  which  steel 
armor  wire  sheathing  is  placed 
to  provide  mechanical  strength 
and  protection.  The  armor  wires 
are  covered  with  inverse  layers 
of  jute  over  which  an  outer 
covering  of  compound  is  applied.  The  overall  diameter  of  deep-sea  cable  varies  from  about 
3/4  in.  to  1  1/2  in.,  depending  on  the  size  of  conductor  and  the  construction  employed. 

Table  1  shows  certain  properties  of  non-loaded  cables  now  in  operation. 

Table  1.    Certain  Properties  of  Non-loaded  Beep-sea  Cables 


PIG.  1.    Non-loaded  Submarine  Cable  Construction 


Weight,  pounds  per 

Diameter 

Diameter 

Resistance, 

Capacitance, 

nautical  mile 

of 

over 

ohms  per 

microfarads 

Copper 

Gutta-percha 

Conductor, 
mils 

Gutta-percha, 
mils 

nautical  mile 
at  75  deg  fahr 

per  nautical 
mile 

70 

120 

70 

252 

16.90 

0.272 

107 

120 

86 

258 

11.05 

0.316 

107 

166 

86 

298 

11.05 

0.280 

130 

130 

95 

270 

9.10 

0.334 

140 

140 

99 

280 

8.45 

0.335 

160 

150 

106 

291 

7.40 

0.345 

180 

160 

112 

302 

6.58 

0.351 

200 

180 

114 

318 

5.92 

0.339 

225 

225 

124 

354 

5.25 

0.332 

275 

225 

138 

360 

4.30 

0,363 

350 

300 

151 

412 

3.38 

0.347 

500 

315 

180 

432 

2.37 

0.398 

650 

400 

203 

487 

1.82 

0.398 

700 

360 

211 

470 

1.69 

0.435 

The  shore  ends  of  ocean  cables  are  usually  of  the  twin-conductor  type,  a  cross-section 
of  which  is  shown  in  Fig.  2.  The  second  conductor  in  the  end  cable  is  used  to  extend 
the  circuit  ground  terminations  out  into  deep  water  for  the  purpose  of  minimizing  extra- 


Deep  Sea 


Conductor 


Shoce  i3rnH 
Dotfele  Yarn  Serving  an 


PeraaaOoyTa 

Copper  Tap«s 


Copper 
Wire 


FIG.  2.     Loaded  Deep-sea  Cable  Construction 


neous  ground  disturbances  from  power  circuits  and  natural  causes.    Since  the  end  sections 
of  cable  usually  rest  in  shallow  water  and  are  subject  to  severe  water  action  and  other 

*  Deproteinized  rubber  has  been  employed  as  a  substitute  for  gutta-percha,  and  a  synthetic 
insulation  of  polyethylene  is  being  experimented  with. 


18-42 


TELEGKAPHY 


disturbances  near  shore,  the  construction  is  much  heavier  than  for  the  deep-sea  sections, 
both  in  armoring  and  jute  layers,  so  that  the  overall  diameter  may  be  as  large  as  4  in. 

Loaded  cables,  in  their  make-up,  closely  resemble  the  non-loaded  cables  just  described, 
the  principal  difference  being  that  the  copper  conductor  or  conductors  (for  twin  conductor) 
are  spirally  wrapped  with  a  thin,  high-permeability  tape  of  Permalloy  (alloy  of  nickel 
and  iron)  (see  Fig.  2) .  This  tape  uniformly  and  continuously  loads  the  copper  conductors 
and  results  in  material  signal  transmission  improvements  (over  the  non-loaded  cable), 
such  as  lower  and  more  uniform  attenuation  with  frequency  and  less  distortion. 

Table  2  shows  certain  properties  of  loaded  cable  now  in  service. 

Table  2.    Certain  Properties  of  Loaded  Deep-sea  Cables 


Weight,  pounds  per 
nautical  mile 

Diameter  in 
Mils-over 

Resistance, 
ohms  per 
nautical  mile 
at  75  deg  fahr 

Capacitance, 
microfarads 
per  nautical 
mile 

Inductance, 
millihenries 
per  nautical 
mile 

Copper 

Gutta- 
percha 

Loading 
Material 

Con- 
ductor 

Load- 
ing 

Gutta- 
percha 

573 

517 
255 

277* 
605  * 

387 
355 
252 
258 
370 

72 
61 
43 
73 
104 

180 
171 
121 
126 
182 

192 
182 
132 
148 
202 

480 
430 
360 
375 

2.09 
2.31 
4.65 
4,28 
1.97 

0.370 
0.375 
0.318 
0.340 
0.393 

63 
86 
140 
170 
118 

*  Approximate  values. 


12.  OPERATION 


Long  non-loaded  submarine  cables  (the  longest  being  about  3500  nautical  miles)  have 
high  values  of  resistance  and  capacitance,  which  attenuate  and  distort  the  telegraph  sig- 
nals to  such  an  extent  that,  until  recently,  the  usual  methods  of  land  line  operation  could 
not  be  employed. 

Early  methods  of  transmission  employed  a  modified  form  of  bridge  duplex  with  artificial 
line,  giving  duplex  operation,  and  electromechanical  types  of  receiving  equipment. 

Signals  were  transmitted  in  the  cable  Morse  Code  from  a  perforated  tape  by  a  trans- 
mitter and  a  group  of  associated  relays,  arranged  to  apply  positive  or  negative  battery 
or  ground  to  the  apex  of  the  bridge  circuit  of  the  duplex  set  in  accordance  with  the  code. 

The  receiving  instrument  first  used  on  transatlantic  cables  was  a  moving-coil-type 
mirror  galvanometer,  connected  as  such  in  the  bridge  duplex  circuit.  Because  of  its 
sensitivity  and  favorable  signal-shaping  characteristics,  this  instrument  was  used  for 
many  years,  until  replaced  by  the  siphon  recorder,  which  recorded  on  a  moving  paper 
tape  in  ink  the  variations  in  magnitude  of  the  incoming  signals.  The  recorder  was  later 
supplemented  by  (1)  various  types  of  magnifiers,  which  amplified  the  received  signals 
before  the  signals  reached  the  recorder,  and  (2)  sensitive  cable  relays,  such  as  the  drum 
and  gold-wire  types,  which  permitted  dispensing  with  manual  relay  operation  at  repeater 
stations. 

These  developments,  together  with  the  application  of  regenerative  repeaters,  extending 
to  the  period  immediately  after  World  War  I,  made  possible  greater  signaling  speeds,  but 
with  recorder  operation  it  was  still  necessary  to  transcribe  messages  manually  from  the 
recorder  tape  at  the  receiving  terminal.  Satisfactory  operation  required  that  the  unbal- 
ance current  in  the  receiving  arm  of  the  bridge  should  not  exceed  one-sixth  the  value  of 
the  received  signaling  currents,  thus  necessitating  the  maintaining  of  a  duplex  balance 
between  the  cable  and  artificial  line  within  about  1/100  per  cent. 

Because  of  the  high  cost  of  submarine  cables,  intensive  study  and  experimental  work 
have  been  continuous,  new  techniques  and  devices  being  sought  for  increasing  the  effi- 
ciency of  these  cables.  The  electromechanical  receiving  equipment  was  necessarily 
fragile  to  respond  to  weak  signals,  and  the  gain  of  the  magnifiers  was  relatively  low  as 
compared  to  electronic  amplifiers.  Attempts  in  1918-1919,  to  apply  vacuum-tube  signal- 
shaping  amplifiers  to  improve  signal  reception  did  not  result  favorably,  mainly  because 
of  (1)  high  level  disturbances  existing  at  that  time  on  duplexed  non-loaded  cables,  due  to 
interference  and  duplex  unbalances,  and  (2)  the  fact  that  suitable  electrical  networks, 
equivalent  to  or  better  than  the  mechanically  tuned  moving  coil,  were  not  available. 

With  the  laying  of  the  first  loaded  oceanic  cable  in  1924,  higher  signaling  cable  speeds 
were  possible,  limited  by  recorder  operation  and  other  terminal  equipment.  Concentrated 
effort  toward  improving  this  equipment  resulted  in  the  development  of  a  signal-shaping 
amplifier  and  a  multiplex  printer  system  suitable  for  high-speed  loaded-cable  operation. 
This  equipment  was  installed  on  the  first  and  subsequent  loaded  cables,  resulting  in  rais- 


OPERATION 


18-43 


ing  the  message  capacity  of  these  cables  as  much  as  three  to  eight  times  over  that  of  the 
older,  non-loaded  systems.  These  loaded  systems  are  still  giving  satisfactory  service. 

The  substantial  gams  made  in  cable  message  transmission  through  development  of  the 
loaded-cable  system  called  forth  increased  effort  toward  bettering  the  non-loaded  cable 
performance.  The  results  so  far  attained  have  been  successfully  applied  quite  extensively 
in  the  North  Atlantic  and  in  the  Alaska  communications  cable  systems,  having  been 
accelerated  by  the  needs  of  World  War  II. 

The  improvement  program  in  non-loaded  submarine  cable  operation  has  included  (1) 
conversion  from  cable  code  recorder  operation  to  five-unit  code  printer  operation,  (2) 
replacement  of  magnifiers  with  vacuum-tube  signal-shaping  amplifiers  which  permit  the 
use  of  rugged  land-line-type  polar 
relays,  and  (3)  improvements  hi 
duplex  artificial  line  networks  and 
in  the  technique  of  balancing. 
Thus,  the  advantages  of  more 
nearly  automatic  operation,  in- 
creased circuit  speeds,  and  reduced 
maintenance  at  cable  stations  have 
been  secured. 

The  printer  system  standardized 
for  non-loaded  cables  is  funda- 
mentally similar  to  the  loaded- 
cable  system  but  more  nearly  re- 
sembles the  land  line  multiplex.  _  ^  _  .  ^ 

T±  ™«l™c.  -,-,0/v  rt-p  •*-'k« +0-^^  4-  ««,-,»«*+  ^IG-  3.  Typical  Arrangement  of  Terminal  tmts  for  Du- 
It  makes  use  of  the  tape  transmit-  plexed  NonlYoaded  Cabfe  (Courtesy  Western  Union  Tde- 
ter,  rotary  distributor,  synehroniz-  graph  Co.,  A.LEJE.,  and  C.  H.  Cramer) 

ing  mechanisms,  and  printers,  thus 

providing  for  integrated  operation  with  land  line  systems.  It  differs  from  these  latter 
systems  in  that  it  is  applied  to  single  as  well  as  multichannel  operation,  whereas  the  land 
line  system  requires  start-stop  seven-unit  code  printers  for  single-channel  use;  and  on 
long  cables  the  printer  signals  are  transmitted  at  a  speed  such  that  pulses  of  unit  or  dot 
length  are  received  at  very  small  amplitude  and,  in  effect,  are  considered  as  absent. 

The  receiving  networks  are  adjusted  to  respond  to  signals  two  or  more  units  long,  so 
that,  from  the  standpoint  of  signal  reception,  the  fundamental  received  frequency  is  one- 
half  the  transmitted  dot  frequency.  The  receiving  relay  operates  on  a  three-position 
basis,  remaining  at  the  zero  position  for  dot  signals,  which  are  reinserted  synchronously 

by  the  receiving  rotary  distributor.  The  at- 
tenuated-dot method  of  transmission,  though 
appearing  to  permit  doubling  the  cable  output 
as  compared  with  normal  multiplex  trans- 
mission, only  approaches  such  output  as  a 
limit.  Actually,  the  increase  in  letters-per- 
minute  circuit  speed  is  only  about  80  per  cent 
over  that  attainable  with,  normal  transmis- 
sion, because,  with  attenuated-dot  transmis- 
sion, the  received  signals  are  more  susceptible 
to  interference  and  more  difficult  to  shape. 
With  this  method,  the  five-unit  code  used  in 
land  line  multiplex  functions  as  a  2.5-unit  code, 
giving  a  net  gain  over  the  3.7-unit  recorder 
Morse  code. 

The    cable    printer    system    is    flexible    in 
that  individual  channels  may  be  terminated, 
extended,   or  combined  with  other  channels 
to  satisfy  traffic  requirements  and  the  trans- 
Channel  efficiency  may  be  increased  by  the  a^ppli- 


Resonant-frequency 
cycles  per  Second 
17.5 
J-      2.16  MH 


FIG.  4.  Typical  Arrangement  of  Resonant 
Balance  Networks  (Courtesy  Western  Union 
Telegraph  Co.,  A.I.E.E.,  and  C.  H.  Cramer) 


mission  speeds  of  the  available  circuits, 
cation  of  land  line  automatic  systems. 

Figure  3  shows  a  typical  arrangement  of  important  terminal  units  for  duplexed  non- 
loaded  cable,  including  a  pre-amplifier  shaping  network,  an  amplifier  tinit,  receiving  relays, 
and  a  local  correction  network.  Resonant  balance  networks,  as  shown  in  Fig.  4,  are  also 
provided  in  a  balanced  arrangement  between  the  series  condensers  and  the  bridge  points  of 
the  receiving  circuit. 

The  pre-amplifier  shaping  network,  shown  diagrammatically  in  Fig.  5,  has  been  designed 
to  meet  certain  requirements  which  permit  satisfactory  printer  operation  on  long  non- 
loaded  cables  at  the  highest  practicable  speeds,  the  more  important  being: 


18-44 


TELEGRAPHY 


Input  to 
amplifier 


1.  A  higher  standard  of  receiving  accuracy,  continuity,  and  reliability  of  operation 
than  with  recorder  operation. 

2.  The  ability  to  equalize  or  restore  the  received  frequency  components  of  the  signals 
to  an  approximation  of  the  original  amplitude  and  phase  relationships. 

3.  Passing  only  the  narrowest  band  of  frequencies,  consistent  with  avoiding  undue 
characteristic  distortion  and  with  limiting  the  effects  of  extraneous  interference  and 
duplex  unbalance  voltages.     Direct  cm-rents  and  alternating  currents  of  near-zero  fre- 
quency must  be  rejected,  in  order 

L°fTite8r?S       resonant  to  avoid  effects  from  earth  currents, 

particularly  as  a  result  of  magnetic 

storms. 

4.  A  shaping  network  electrically 
symmetrical    with   respect   to   the 
duplex  bridge  or  electrically  isolated 
from  the  bridge. 

5.  Network  elements  designed  for 
a  wide  range  of  adjustment. 

6.  Amplifier  gain  and  output  suf- 
ficient to  operate  rugged  polar  re- 
lays, similar  to  those  used  in  land 
line  systems. 

This  pre-amphfier  network,  when 
P^rly  adjusted  passes  a  band 
of  frequencies,  including  those  com- 
ponents required  for  good  signal  shape,  properly  proportioned  and  phased,  and  largely 
or  almost  completely  suppresses  those  frequencies  below  the  fundamental  received  fre- 
quency. The  first  branch  of  the  network,  being  symmetrical  and  directly  across  the 
duplex  bridge,  is  tuned  to  about  1.5  times  the  received  frequency.  The  remaining  ele- 
ments of  the  network,  being  isolated  from  the  bridge  by  shielded  transformers,  are 
arranged  in  two  paths  which  combine  at  the  amplifier  input. 

The  lower  frequencies  pass  to  the  amplifier  over  the  low-frequency  path  with  little,  if 
any,  further  shaping,  while  the  higher  frequencies  pass  to  the  amplifier  over  the  high- 
frequency  path,  in  which  they  are  further  shaped  by  a  bridge-type  phase-adjusting  net- 
work, a  low-pass  filter  for  added  suppression  above  the  required  signal  band,  a  parallel 
resonant  circuit  tuned  to  1.5  times 
the  received  frequency,  and  suit- 
able resistance  controls.  The 
phase-adjustment  network  pro- 
vides the  required  wave-front 
steepness  with  less  higher-fre- 
quency components,  thus  permit- 
ting further  discrimination  against 
unwanted  frequencies. 

The  amplifier  unit  is  of  the 
three-stage  resistance-capacitance 
coupled,  pushpull  type  with  two 
stages  of  voltage  amplification. 
Frequencies  below  those  received 
from  the  shaping  network  are  sup- 
pressed. The  maximum  voltage 
gains  and  available  overall  gains 
are  83  db  and  103  db,  respectively. 
The  amplifier  output  is  adequate 


B 


Tims 


FIG.  6.  Received  Signal  Resulting  from  Transmission  of 
Long  Signal  over  Non-loaded  Submarine  Cable  (Courtesy 
Western  Union  Telegraph  Co.,  A.IJE.E.,  and  C.  H.  Cramer) 


A.  Component  through  high-frequency  path  of  Fig.  5. 
5.  Component  through  low-frequency  path  of  Fig.  5. 

C.  A  +  B. 

D.  Component  supplied  by  local  correction  network. 
-    .        , ,      ,    ,      ,   ,      j  , .               E.  Complete  signal,  C  +  D. 

for  operating  the  latest  land-line- 
type  polar  relays. 

Two  standard  two-position  re- 
lays function  in  unison  as  a  three- 
position  relay,  in  accordance  with  usual  cable  practice,  but  the  circuit  is  readily  converti- 
ble, if  desired,  to  receive  two-current  signals. 

Figure  6  shows  the  shapes  of  the  component  signals  in  the  formative  stages  and  the 
shape  of  the  complete  signal  at  the  amplifier  input.  Curve  C  shows  the  signal  shape  as  it 
leaves  the  pre-amplifier  network,  at  which  point  signals  of  the  fundamental  received  fre- 
quency are  fully  shaped  but  longer  signals  would  be  badly  distorted  because  of  lack  of 
low-frequency  components.  The  local  correction  network  (Fig.  3)  functions  to  restore 
these  components  under  control  of  the  receiving  relays.  The  shaped  local  correction 


OPEEATION 


18-45 


voltages,  curve  Z>,  are  added  to  the  received  signal  in  the  grid  circuit  of  the  output  stage 
of  the  amplifier,  resulting  in  the  fully  shaped  and  complete  signal,  curve  E. 

For  loaded-cable  systems,  essentially  all  signal  shaping  occurs  in  the  pre-amplifier  and 
interstage  networks,  but  there  is  some  deficiency  in  the  very  low-frequency  components. 
The  method  of  shaping  employed  with  the  non-loaded  cable  systems  not  only  affords 
greater  immunity  from  low-frequency  disturbances  but  simplifies  amplifier  design  and 
stability  and  eliminates  slow  transients  (wandering  zeros)  . 

Duplex  operation  of  non-loaded  cables,  giving  greater  total  message  capacity,  is  nor- 
mally used.  With  the  advent  of  modern  amplifiers,  extraneous  interference  and  duplex 
unbalance  levels  limit  signaling  speeds.  If  un- 
balance is  governing,  maximum  capacity  is  ob- 
tained by  using  unequal  speeds  in  the  two  direc- 
tions of  transmission,  but  duplex  unbalance  is 
at  present  less  of  a  factor,  owing  to  improved 
artificial  lines  and  balancing  methods. 

The  basic  ocean  cable  artificial  line  is  still 
about  the  same  as  it  was  at  the  beginning  of 
duplex  operation.  The  lumped  series  resist- 
ances and  shunt  capacitances,  simulating  cor- 
responding cable  constants,  have  been  subdi- 
vided and  arranged  for  greater  flexibility  of 
adjustment.  Modern  artificial  lines  of  Ameri- 
can design  are  subdivided  so  that  the  lumped 
values  of  resistance  and  capacity  increase  pro- 
gressively as  their  distance  from  the  head 
(point  nearest  the  apex)  of  the  artificial  line 
increases. 

The  cable  circuit  parameters  include  induct- 
ance and  effective  resistance,  which  vary  with 
frequency  because  of  the  earth-return  path 
characteristics,  and  are  known  as  the  sea-return 
impedance.  These  factors,  though  small,  are 
important  in  high-accuracy  balancing  of  the 
near  end  of  the  cable.  By  using  tapered  re- 
sistance values  in  series  with  the  shunt  capac- 
itance elements  of  the  artificial  line  sections, 
the  sea-return  impedance  can  be  simulated 
over  a  relatively  wide  frequency  band  and  the 
propagation  constant  of  the  sections  to  which 
these  resistances  are  added  is  not  materially 
changed. 

However,  in  order  to  meet  balance  require- 
ments, further  refinements  in  duplex  balanc- 
ing are  necessary.  Slow  reversals  are  trans- 
mitted, and  observations  of  the  residual  unbal- 


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Frequency-cycles  per  second 

A.  Before  insertion  of  networks  of  Fig.  4. 

B.  After  insertion  of  networks  of  Fig.  4. 

FIG.  7.    Frequency  Characteristic  of  Unbalance 

on  a  Transatlantic  Cable    (Courtesy  Western 

Union   Telegraph   Co.,   A.I.E.E.,  and  C.   H. 

Cramer) 


ance  transient  are  made  with  a  cathode-ray  oscillograph  or  ink  recorder.  In  the  final 
adjustments,  unbalance  at  operating  speeds  is  also  observed.  The  artificial  line  adjust- 
ments have  their  limitations  (which  are  not  usually  the  same  for  different  cables),  because 
of  inadequacy  of  artificial  line,  interferences,  or  other  factors. 

Corrective  resonant  networks  are  designed  to  secure  the  ultimate  duplex  balance  needed 
for  satisfactory  operation.  These  networks,  Fig.  4,  supplement  the  artificial  line  and 
provide  an  impedance,  which  can  be  effectively  controlled  as  to  magnitude  and  width  of 
frequency  band.  The  networks  are  designed  in  accordance  with  a  frequency  character- 
istic of  the  residual  unbalance. 

This  frequency  characteristic  is  obtained  by  taking  electrical  measurements,  directly 
across  the  duplex  bridge  over  the  important  frequency  range,  of  the  effective  residual 
impedance  unbalance  at  the  head  of  the  artificial  line.  Several  of  the  single  networks  are 
usually  used,  some  of  them  being  inserted  in  series  with  the  cable  and  some  of  them  in 
series  with  the  artificial  line  and  adjusted,  as  required.  Final  adjustment  is  made  in  the 
usual  artificial  line  controls. 

Figure  7  shows  the  frequency  characteristic  of  unbalance  on  a  transatlantic  cable  before 
and  after  insertion  of  the  corrective  resonant  networks  and  the  final  adjustments.  The 
balance  has  been  improved  by  these  networks  over  the  most  important  frequency  band, 
with  the  greatest  improvement  at  the  higher  frequencies. 


18-46  TELEGRAPHY 

Crossfire  between  cables  usually  occurs  if  two  or  more  cables  land  at  the  same  point  and 
extend  underground  to  the  cable  station  with  small  separation.  This  interference  is  con- 
trolled by  applying  simple  corrective  networks  at  the  cable  station.  If  two  cables  land 
at  different  points  and  are  laid  closely  parallel  for  some  distance,  the  corrective  problem  is 
more  difficult,  requiring  more  complex  networks. 

With  the  techniques  that  have  been  developed  for  improved  duplex  balances  and  the 
correction  of  crossfire,  extraneous  interference,  mostly  of  natural  origin,  becomes  con- 
trolling in  signal  speed.  Natural  interference  is  picked  up  in  the  shallow-water  end  sec- 
tions of  cables,  its  magnitude  varying  widely  from  cable  to  cable,  depending  on  the  depths 
and  distances  encountered.  Receiving  earths  of  the  non-loaded  cables  are  usually  located 
up  to  several  miles  from  shore.  The  twin-conductor  end  sections  of  cable  greatly  assist  in 
limiting  the  interference  and  could  be  extended  further  into  the  ocean,  but  the  cost  would 
be  correspondingly  increased. 

Some  improvement  in  interference  levels  has  been  obtained  by  increasing  the  sending 
voltages,  restricted  until  recent  years  to  about  50  volts,  up  to  90  to  120  volts,  now  in  com- 
mon use  by  some  of  the  cable  companies. 

Signal-shaping  amplifiers  of  the  type  just  described  are  being  used  on  many  of  the  longer 
transoceanic  cables;  short-cable  amplifiers  are  employed  for  short  connecting  cables. 

On  certain  duplex-operated  non-loaded  cables,  the  sum  of  the  letters-per-minute  speeds 
in  the  two  directions  now  averages  in  printer  operation  over  40  per  cent  more  than  with 
the  previous  recorder  speeds.  The  average  net  increase  in  the  message  capacity  of  these 
cables,  considering  short-cut  methods  permissible  in  recorder  but  not  printer  operation,  is 
over  30  per  cent.  In  one  case,  it  is  expected  that  the  ultimate  speed  in  printer  operation 
when  three-channel  multiplex  equipment  is  available  will  be  750  letters  per  minute  in 
each  direction  of  transmission. 

In  loaded  cables,  the  channel  speed  is  usually  250  to  300  letters  per  minute,  the  number 
of  channels  per  cable  varying  between  4  and  8,  depending  on  the  cable  make-up.  One 
loaded  cable,  about  2300  nautical  miles  long,  operates  at  65  cycles  per  second  and  provides 
5  one-way  channels,  each  with  a  capacity  of  312  letters  per  minute. 

Power  equipment  consists  of  storage  batteries  and  charging-equipment  installations  of 
the  capacities  and  sizes  necessary  to  provide  both  regular  and  emergency  power  at  the 
cable  stations.  The  d-c  voltages  range  from  90  to  120  volts. 


TELEGRAPH  EQUIPMENT 

By  John  D.  Taylor 

The  telegraph  central  office  is  the  centralizing  point  which,  for  its  particular  area, 
directs  and  regulates  the  movement  of  telegraph  messages.  From  such  points,  the  operat- 
ing personnel  also  maintains  constant  supervision  over  the  proper  functioning  of  and 
needed  repairs  to  equipment  and  outside  plant. 

13.  CENTRAL-OFFICE  EQUIPMENT 

Central-office  equipment  for  telegraph  operation  consists  of  a  wide  variety  of  equip- 
ment units  and  associated  facilities.  The  types  and  amounts  of  equipment  at  any  given 
central  office  vary  over  a  wide  range,  depending  mainly  upon  its  importance  in  the  general 
telegraph  network,  the  message  volume  handled,  and  the  nature  of  the  traffic.  Some  of 
this  equipment  has  previously  been  described  and  will  not  be  discussed  further  here. 

In  a  large  office,  the  principal  classifications  of  equipment  may  be  considered  to  include: 

1.  Terminal  equipment  for  open-wire  lines — entrance  cables,  protector  and  distributing 
frames,  and  test  board  (Western  Union  designation  is  line-terminal  switchboard) . 

2.  Intermediate  operating  equipment — repeaters,  concentrators,  and  system  apparatus, 
such  as  multiplex,  varioplex,  reperforator-switching,  carrier,  and  Telefax. 

3.  Operating  positions — teletypewriter  switchboard  (Bell  System  designation),  operat- 
ing tables  or  positions  for  Morse,  multiplex,  and  teleprinter  sending  and  receiving  appa- 
ratus, tape  perforators,  reperforators  and  transmitters,  Telefax  terminals,  telephone  lines, 
and  switching  facilities. 

4.  Message-handling  equipment — belt  conveyors  between  incoming  and  outgoing  operat- 
ing positions  and  distributing  centers  within  an  office,  and  pneumatic-tube  terminals 
from  branch  or  other  message-handling  centers  to  the  main  office. 


CENTRAL-OFFICE  EQUIPMENT  18-47 

5.  Power  equipment — power-generating  equipment,  such  as  rectifiers  and  motor-genera- 
tors, power  switchboards,  power  distributing  systems  to  power-operated  units,  storage 
batteries,  and  emergency  power  plants. 

6.  Building    equipment — lighting,    heating,    ventilation,    elevator    service,    personnel 
quarters,  and  many  other  items  of  this  nature. 

The  equipment  layout  in  an  office  should  provide  a  minimum  travel  time  of  the  operat- 
ing personnel  in  their  regular  work  and  should  limit  wiring  and  cabling  requirements  be- 
tween equipment  units. 

Protector  (main)  frames  provide  for  termination  of  the  entrance  cables  through  which 
the  open-wire  lines  extend  into  the  central  office,  either  directly  on  protectors  or  on  ter- 
minal blocks,  from  which  the  lines  are  connected  to  protectors.  The  protectors,  consisting 
of  heat  coils  (or  fuses)  and  carbon  block  discharge  gaps,  function  to  prevent  excessive  for- 
eign currents  or  voltages  from  damaging  the  central-office  cables  and  equipment,  as  dis- 
cussed in  more  detail  in  Sections  10  and  17.  Office  circuit  fuses  are  also  provided  to  pre- 
vent excessive  office  currents  from  damaging  the  equipment  or  wiring. 

Testboards  or  line-terminal  switchboards,  to  which  the  lines  are  extended  from  the 
protector  frames  in  cable,  are  designed  to  terminate  the  telegraph  circuits,  entering  an 
office,  in  jacks  for  testing,  patching,  and  other  purposes,  as  required  in  maintaining  and 
operating  these  circuits.  Certain  central-office  equipment  units,  battery  taps,  and  special- 
purpose  apparatus  are  also  terminated  at  such  boards  and  may  be  associated  with  or  dis- 
connected from  the  various  circuits,  to  meet  operating  needs.  In  the  older-type  boards 
telegraph-circuit  layouts  are  usually  established  in  part  by  means  of  patching  cords;  in 
the  latest-type  boards,  these  circuits  are  wired  through  groups  of  jacks,  individual  to  each 
circuit,  eliminating  the  need  for  patching  cords,  except  for  testing  or  establishing  other 
than  the  normal  circuit  layout. 

Intermediate  distributing  frames  have  mounted,  on  their  two  sides,  terminal  blocks, 
to  which  the  various  equipment  units  in  the  office  are  wired.  Office  cables  also  extend 
from  these  frames  to  testboards  and  line-terminal  switchboards  and  to  telegraph  operat- 
ing positions  and  teletypewriter  switchboards,  so  that  by  means  of  crossconnections  on 
these  frames  circuits  may  be  connected  to  the  various  test  board  and  switchboard  jacks, 
operating  positions,  and  equipment  units,  as  desired. 

Teletypewriter  switchboards  employed  for  the  purpose  of  establishing  connections  be- 
tween teletypewriter  subscribers  consist  principally  of  positions  equipped  with  jacks,  cord 
circuits,  and  a  keyboard  sending  and  receiving  teletypewriter,  which  may  be  associated 
by  means  of  keys  with  any  cord  circuit  on  the  position. 

The  operator  handles  connections  somewhat  like  an  operator  at  a  manual  telephone 
switchboard  position,  the  principal  difference  being  that  the  incoming  calls  are  answered 
and  extended  to  the  called  subscriber  by  operating  the  teletypewriter,  with  the  assistance 
of  similarly  equipped  distant  operators,  if  necessary. 

These  boards  are  designed  to  serve  as  few  as  10  lines  (mostly  for  private  networks)  or  as 
many  as  2040  subscriber  lines  and  600  intertoll  trunks,  when  the  outward,  inward,  and 
through  traffic  is  handled  at  one  board. 

In  order  to  improve  transmission  from  a  central  office  to  a  teletypewriter  station,  a 
wave-shaping  network,  consisting  of  resistance,  inductance,  and  capacitance  of  various 
values  and  combinations,  depending  on  the  type  of  loop  and  connected  equipment,  is  fre- 
quently inserted  in  the  side  of  the  loop  connected  to  the  repeater  at  the  central  office. 
Wave-shaping  networks  are  also  employed,  as  required,  at  the  stations.  These  networks 
assist  in  restoring  the  received  signal  wave  to  its  original  shape. 

The  jacks,  cords,  and  plugs  used  at  telegraph  switchboards  and  testboards  for  testing, 
patching,  and  establishing  connections  may  be  of  standard  types,  such  as  those  in  manual 
telephone  testboards  and  switchboards.  However,  where  low-resistance  conductors  with 
greater  service  margins  are  needed,  these  units  are  frequently  of  heavier  construction. 
The  number  of  conductors  will  vary  between  different  boards,  depending  on  the  circuit 
requirements. 

Telegraph  repeaters  employing  polar  transmission  are  standard  for  d-c  trunk-line 
terminal  sets;  they  usually  operate  on  a  full  duplex  basis  with  ground  return.  Other  types 
of  repeaters  are  used  at  intermediate  trunk-line  points  for  transmission  reasons.  The 
operating  functions  of  repeaters  have  been  discussed  in  article  6. 

Repeaters  are  used  in  various  arrangements  in  circuits,  the  name  by  which  they  are 
designated  indicating  the  manner  in  which  they  function,  such  as  a  combination  duplex- 
duplex  half  repeater,  terminal  duplex-duplex  half  repeater,  and  high-speed  polar  duplex, 
high-speed  single-line,  and  regenerative  repeater.  One  form  of  repeater  provides  for 
receiving,  recording,  and,  when  the  outgoing  circuit  is  clear,  retransmitting  telegraph  sig- 
nals, which,  in  effect,  is  equivalent  to  storing  signals.  Special  types  of  repeaters  serve 
other  purposes,  such  as  connecting  multiplex  channels  to  other  channels  and  loops. 


18-48 


TELEGRAPHY 


Relays  perform  vital  functions  in  the  operation  of  telegraph  circuits  and  equipment. 
The  modern  high-speed  polar  relay  operates  efficiently  and  with  precision.  Figure  1 

shows  a  common  type  of  polar  relay  for 
use  in  high-speed  d-c  telegraph  circuits. 
Figure  2  shows  a  plug-type  polar  relay, 
commonly  used  in  d-c  telegraph  line  cir- 
cuits. Many  other  types  of  relays  are  em- 
ployed in  telegraph  circuits,  each  designed 
to  perform  its  particular  function. 

Multiplex  distributors,  as  previously  ex- 
plained in  this  section,  are  devices  having 
segmented  and  solid  ring  face  plates,  with 
which  rotating  brushes  are  in  contact,  and 
by  means  of  which  telegraph  signals  from 
one  or  more  circuits  are  transmitted  over 
a  single  telegraph  line  in  sequence  on  a 
time-sharing  basis.  The  face  plates  are 
removable  and  may  be  changed  as  desired 
to  meet  operating  requirements. 

The  brushes  are  mounted  on  a  shaft, 
driven  by  an  impulse  motor,  which  is  syn- 
chronized with  the  motor  of  a  multiplex 
set  at  the  distant  line  terminal.  For  this 
reason,  the  start  and  stop  pulses  are  not 
required,  as  they  are  for  the  teleprinter  or 
teletypewriter,  and  only  a  five-unit  code  is 
employed  per  character  for  each  circuit 
operating  over  the  multiplex  line. 

The  shaft  between  the  motor  and  the 
brush  assembly  consists  of  two  parts  joined 
together  by  a  magnetically  operated  ratchet 
device,  by  means  of  which  the  angular  posi- 


FIQ.  1.    Typical  High-speed  D-c  Telegraph  Polar 

Relay  (Courtesy  Western  Union  Telegraph  Co., 

A.I.E.E.,  F.  E.  D'Humy,  and  P.  J.  Howe) 


tion  of  the  brushes  with  respect  to  the  motor  rotor  may  be  changed  in  steps  of  1 1/2  angular 
degrees.  The  change  may  be  made  with  the  motor  operating,  if  desired.  A  mercury- 
filled  flywheel  mounted  on  the  motor  shaft 
provides  stability  of  rotation.  The  various 
connections  to  the  distributor  are  brought 
out  to  multicontact  bayonet-type  plugs 
to  provide  for  rapid  replacement  of  the 
distributor  in  case  of  trouble.  Figure  3 
shows  one  type  of  multiplex  distributor. 

The  shaft  speed  is  determined  by  the 
required  channel  speed.  For  four-channel 
operation  in  each  direction,  at  a  channel 
speed  of  66  words  per  minute,  the  shaft 
speed  is  about  396  rpm.  However,  the 
total  message  capacity  for  the  line  is  528 
words  per  minute. 

Synchronization  of  speed  between  two 
multiplex  sets  at  opposite  ends  of  a  mul- 
tiplex circuit  is  accomplished  by  means 
of  a  driving  fork  associated  with  each  set. 
This  fork  is  magnetically  vibrated  and 
equipped  with  contacts  to  generate  im- 
pulses from  a  d-c  supply  for  operating  the 
motor  of  the  set.  The  frequency  of  vibra- 
tion of  the  fork,  and  hence  the  motor 


FIG.  2.    Typical  D-c  Telegraph  Relay  for  Line  and 

Other  Circuits  (Courtesy  Western  Union  Telegraph 

Co.,  A.I.E.E.,  F.  E.  D'Humy,  and  P.  J.  Howe) 


speed,  may  be  altered  by  changing  the 
position  of  weights  clamped  to  the  fork 
tines.  The  normal  fork  frequency  (with- 
out weights)  is  about  60  cycles  per  second, 
corresponding  to  a  distributor  speed  of  about  360  rpm.  Forks  with  shorter  tines  are  used 
for  high-speed  circuits.  Figure  4  shows  a  drawing  representative  of  a  driving  fork. 

The  tape  perforator  consists  of  a  perforating  mechanism,  actuated  by  a  keyboard  unit, 
in  which  each  individual  key  lever,  with  certain  exceptions,  is  designated  with  an  upper- 


CENTRAL-OFFICE  EQUIPMENT 


18-49 


and  lower-case  character.  By  operating  the  key  levers  when  the  perforator  is  in  its  operat- 
ing condition,  the  characters  corresponding  to  the  keys  operated  will  be  punched  in  a 
paper  tape  in  the  standard  five-unit  code.  *--~~ 


FIG.  3.    Typical  Rotary  Distributor  (Courtesy  Western  Union  Telegraph  Co.,  A.I.E.E.,  F.  E.  D'Humy, 

and  P.  J.  Howe) 

In  one  type  of  perforator  design,  a  punch  block  contains  six  small  cylindrical  metallic 
fingers  or  punches,  between  the  die  plates  of  which  block  the  tape  is  fed  (see  Fig.  5) .  A 
punch  hammer,  operated  by  a  magnet,  forces  the  punches  through  the  tape  as  it  passes 
the  punch  holes  in  the  die  plates.  As  each  character  is  punched,  the  tape  is  moved  for- 
ward one  space  by  a  pawl  and  feed  roll,  and  the  succeeding  character  is  then  punched. 
Five  of  the  punches  are  for  code  perforations,  and  the  sixth  punch  provides  the  feed  holes 
in  the  center  of  the  tape. 

Between  the  punch  hammer  and  the  five  punches  are  five  punch  bars,  which  are  con- 
nected by  bell  cranks  to  five  U-shaped  bars  (loops),  pivoted  at  each  end  and  held  by 


FIG.  4.     Driving  Fork  Used  in  Multiplex  Systems 

means  of  springs  so  that  their  greatest  length  is  in  a  horizontal  position  directly  beneath 
the  keyboard. 

Attached  to  the  lower  edge  of  each  key  lever  is  a  piece  of  metal,  called  a  comb,  which  is 
cut  out,  so  that  depressing  a  key  will  cause  its  comb  to  strike  the  top  edge  of  one  or  more 
of  the  loops  and  move  them  downward.  The  combs  are  cut  differently  for  the  different 
keys,  resulting  in  a  different  combination  for  each  key  depressed. 


18-50 


TELEGRAPHY 


The  depression,  of  any  loop  moves  the  corresponding  punch  bar  from  in  front  of  its 
punch  so  that,  when  the  punch  hammer  is  operated  by  the  magnet,  the  corresponding 
punch  does  not  operate  and  the  tape  is  not  perforated  by  that  punch.  A  sixth  (power) 


FIG.  5.     Perforator  Punching  Mechanism  (Courtesy  Bell  System) 

loop,  operated  when  any  key  is  depressed,  energizes  the  punch  magnet,  which  actuates 
the  punch  hammer. 

This  perforator  also  provides  for  moving  the  tape  backward  for  correction  of  errors  and 
for  indicating  end  of  a  line,  so  that  the  carriage  return  key  can  be  operated  to  start  a  new 
line  at  the  distant  receiving  machine. 

The  typing  reperforator  is  a  device  for  receiving  messages  from  a  telegraph  circuit  or 
transmitter  and  recording  them  in  tape  by  five-unit  code  perforations  and  by  printing  the 
character  on  the  tape  above  the  corresponding  perforations. 

The  transmitter  and  transmitter-distributor  are  devices  for  translating  code  perforations 
in  tape  into  electrical  impulses,  which  are  transmitted  over  a  connecting  medium  to  a  re- 
ceiving device  for  interpretation  as  signals  of  intelligence.  The  perforations  may  be  in  a 
five-unit,  six-unit,  or  other  code,  depending  on  the  particular  circuit  requirements. 


pper  (spacing)  contacts 


Start  segmen 

_                    Contact  tongues 
III S-topsegm  e  nt- 


Line  battery 


(marking)  contacts 
Distributor  brus! 
Commutator 


FIG.  6.   Diagram  Showing  Transmitter  Contacts  Wired  to  Distributor  Segments  (Courtesy  Bell  System) 


In  the  transmitter-distributor  the  tape  transmitter  establishes  the  code  combinations  to 
be  transmitted,  and  the  commutator  distributor  sends  out  these  combinations  over  the 
line,  as  marking  and  spacing  impulses,  in  their  proper  sequence  and  at  the  desired  speed. 
Both  units  are  driven  by  the  same  speed-regulated  motor. 

For  one  type  of  five-unit  code  transmitter-distributor,  the  five  contact  tongues  (see  dia- 
gram hi  Fig.  6)  of  the  transmitter  move  between  two  sets  of  contacts,  one  set  marking  and 


STATION  EQUIPMENT 


18-51 


the  other  set  spacing.  These  tongues  and  the  multipled  marking  and  spacing  contacts  are 
connected  to  distributor  segments.  In  "make-break"  operation,  battery  is  connected  to 
the  marking  contacts  only. 

The  tongues  are  mechanically  connected  to  the  ends  of  five  pivoted  contact  levers,  each 
of  which  has  three  extensions  A,  B,  and  C,  as  shown  in  Fig.  7.  In  the  unoperated  position 
of  the  contact  lever  bail  (the  position  shown  in  Fig.  7) ,  the  contact  lever  springs  pull  down 
on  the  A  extensions,  causing  the  tape  pins  in  the  C  extensions  to  press  up  against  the 
tape  but  the  upper  contacts  remain  closed.  Since  the  tape  pins  are  spaced  the  same 
distance  apart  as  the  tape  perforations,  any  pin  will  then  pass  through  the  tape  if  there  is 
a  perforation  in  the  tape  above  it.  When  a  pin  moves  through  a  perforation,  the  A  exten- 
sion is  permitted  to  move  down  slightly  under  action  of  its  spring,  thus  opening  its  spac- 
ing and  closing  its  marking  contact  by  the  movement  of  its  contact  tongue.  Where  there 
is  no  perforation  in  the  tape  above  a  pin,  the  pin  is  held  in  its  normal  position  against  the 

Upper  (spacing^ 
contact  screw"  ~^ 


Terminal 


Note:  Tape  pta  shown 
extending  upward  through 
perforation  In  tape. 


-Operating  lever 


FIG.  7.     Tape  Transmitter  Mechanism  (Courtesy  Bell  System) 

tape  and  its  contact  tongue  remains  on  spacing.  Thus,  the  code  perforations  in  the  tape 
determine  the  setting  of  each  of  the  contact  tongues  either  to  marking  or  spacing,  and 
hence  the  polarity  of  the  distributor  segments  connected  to  the  tongues. 

After  each  character  is  transmitted,  the  contact  tongues  are  reset  to  spacing  by  opera- 
tion of  the  operating  lever  and  contact  lever  bail.  This  bail  moves  the  B  extensions  to 
the  left,  withdrawing  the  tape  pins  to  a  position  below  the  tape  guide  surface,  thus  mov- 
ing the  contact  tongues  upward.  Also,  after  each  character  is  transmitted,  a  sixth  (feed) 
lever  is  actuated,  causing  the  tape  feed  mechanism  to  move  the  tape  forward  a  distance 
equal  to  that  between  the  character  punches  in  the  tape. 

An  automatic  stop  is  mounted  on  the  transmitter-distributor  base  to  stop  the  trans- 
mitter if  the  associated  perforator  operation  is  interrupted  or  if  its  speed  becomes  less 
than  that  of  the  transmitter.  This  avoids  tape  mutilation  by  the  transmitter.  The  stop 
consists  of  a  light  metal  lever,  suspended  over  the  tape  loop  between  perforator  and  trans- 
mitter, which  is  raised  when  the  loop  becomes  tight,  opening  the  control  circuit  of  the 
transmitter. 

Message  conveyors  are  employed  in  the  larger  telegraph  central  offices  to  reduce  the 
travel  and  handling  time  for  messages  that  must  be  transported  from  one  location  to 
another  in  the  same  office  or  building.  In  the  largest  centers,  the  total  number  of  mes- 
sages handled  daily  may  average  300,000  or  more. 

Pneumatic  tubes  provide  a  rapid  and  efficient  means  of  transporting  the  original  copies 
of  messages,  being  commonly  employed  between  branch  and  central  offices  and  for  intra- 
departmental  use  in  large  central-office  buildings. 


14.  STATION  EQUIPMENT 

Station  equipment  intended  for  telegraph  purposes  is  of  various  types  and  designs  to 
meet  the  needs  of  the  customer.  A  number  of  the  equipment  units  are  also  applicable  for 
use  at  customer  premises,  such  as  the  teletypewriter,  teleprinter,  perforator,  typing  re- 


18-52  TELEGRAPHY 

perforator,  transmitter-distributor,  Telefax,  ticker,  clocks,  and  various  other  units.  The 
station  equipments  discussed  in  the  following  paragraphs  are  additional  to  those  previ- 
ously discussed. 

Printers.  Two  types  of  teletypewriters  of  the  start-stop,  five-unit  code  type  have  not 
been  previously  described.  One  is  a  motor-driven,  single-magnet,  fixed  paper  carriage, 
typebar,  page  printing  type,  operating  normally  at  speeds  of  240  to  368  operations  (40  to 
60  words)  per  minute.  The  paper  may  be  in  single  sheet  rolls,  usually  8  or  8  1/2  in.  wide, 
or  two  or  more  carbon  copies  may  be  made  by  using  the  proper  paper  assemblies. 

The  other  is  a  teletypewriter,  used  where  not  more  than  one  carbon  copy  is  required 
and  where  a  smaller  machine  than  the  first  is  desired  by  the  customer.  This  machine  is 
a  motor-driven,  single-magnet,  moving  paper  carriage,  typewheel,  page  printing  type, 
operating  normally  at  368  operations  per  minute.  It  uses  paper  8  1/2  in.  wide,  which  may 
be  multiple  wound  for  one  carbon  copy. 

Radio-interference-suppression  apparatus  (filters),  consisting  of  inductance,  capaci- 
tance, and  resistance,  are  employed  in  station  equipment,  usually  in  a  parallel-series  rela- 
tion, across  various  make-break  contacts  in  the  teletypewriter  or  teleprinter.  One  type 
of  suppressor  reduces  induction  at  broadcasting  frequencies,  and  another  type  provides 
suppression  at  both  broadcasting  and  higher  frequencies. 

Selectors  are  used  on  important  circuits  to  provide  a  convenient  means  for  calling 
attendants  at  repeater  or  terminal  stations  to  the  circuit  when  trouble  develops.  They 
are  also  used  on  Morse  wires  and  concentrators  to  enable  one  station  to  call  another  with- 
out calling  in  all  the  other  stations  on  the  same  circuit.  The  selector,  Fig.  8,  contains  a 


FIG.  8.     Typical  Telegraph  Selector 

magnet,  normally  connected  with  the  line  or  line  relay,  which  controls  a  mechanism 
arranged  to  close  a  set  of  local  contacts  only  when  the  magnet  is  operated  by  the  particular 
combination  of  impulses  for  which  the  selector  mechanism  is  adjusted.  The  local  contacts 
of  the  selector  may  be  used  to  operate  either  visual  or  audible  signals  or  to  place  the  local 
receiving  apparatus  in  an  operative  condition.  Signal  combinations  for  operating  the 
selectors  may  be  transmitted  manually  with  a  Morse  key,  or  special  clockwork-driven 
calling  keys  may  be  used  for  the  purpose. 

In  Morse  telegraphy  manual  operation  is  employed,  but,  owing  to  the  growth  of  auto- 
matic transmission  of  messages,  Morse  operation  is  confined  mainly  to  occasional  local 
services  over  short-haul  telegraph  facilities,  such  as  those  to  sporting-event  locations, 
railroad-station  offices,  and  small  towns. 

Morse  circuits  are  operated  either  single  or  duplex,  as  conditions  may  require.  Where 
several  such  circuits  terminate  at  a  central  point,  they  are  usually  connected  into  a  con- 
centrating unit  by  means  of  which  one  operator  can  handle  all  the  circuits  so  concen- 
trated. • 

Figure  9  shows  sketches  of  a  Morse  sending  key,  sounder,  and  relay. 

The  sending  key  has  two  contacts,  normally  held  open  by  a  spring,  which  are  closed 
by  manually  depressing  a  key  lever,  which  also  depresses  the  spring.  When  the  key  is 
released,  the  spring  causes  the  contacts  to  open.  A  second  lever  operates  horizontally  to 
close  the  circuit  at  the  key  when  the  key  is  not  in  use. 

The  sounder  consists  of  an  electromagnet  and  an  armature  which  moves  a  sounding 
lever  between  two  adjusting  stop  screws,  the  assembly  being  mounted  on  a  sound-amplify- 


TRANSMISSION  STANDARDS 


18-53 


Morse  Sending  Key 


ing  base.    When  the  magnet  is  energized,  the  armature  is  drawn  down  until  its  stop  screw 
contacts  the  metal  frame,  producing  an  audible  click.    When  the  magnet  is  de-energized, 
the   armature  is  restored  to  its  unoperated  posi- 
tion by  spring  action,  its  outer  end  striking  the 
upper  stop  screw,  which  produces  a  click. 

Two  of  these  clicks  separated  by  a  short  inter- 
val are  interpreted  by  the  operator  as  a  dot.  For 
a  longer  interval  between  clicks  (usually  three  times 
the  interval  between  dots),  the  signal  is  interpreted 
as  a  dash.  The  armature  travel  and  restoring 
spring  tension  can  be  adjusted  to  suit  the  operator. 

Local  sounders  operate  in  local  sounder  circuits; 
they  may  be  of  low  resistance  (about  4  ohms), 
requiring  an  operating  current  of  about  250  ma, 
or  of  high  resistance  (100  or  400  ohms),  requir- 
ing operating  currents  of  f  about  60  and  30  ma, 
respectively.  Main  line  sounders  are  designed 
for  operation  in  series  with  the  main  line  cir- 
cuit. These  sounders  may  be  adjusted  so  that 
their  operation  is  not  materially  affected  by  line 
leakage,  usually  encountered,  and  their  resistance 
may  be  30,  100,  or  120  ohms,  depending  on  circuit 
requirements. 

The  relay  consists  of  an  electromagnet  having  one 
or  more  windings  arranged  to  move  an  armature, 
which  operates  between  a  set  of  contacts.  Signal- 
ing impulses,  passing  through  the  operating  wind- 
ing of  the  relay,  cause  the  armature  to  move  to 
its  proper  contact,  in  accordance  with  the  type 
of  impulse  received,  thus  repeating  the  sent  signals 
in  a  local  circuit  connected  to  the  relay  contacts. 

The  type  of  relay  shown  in  Fig.  9  usually  is 
equipped  with  a  magnet  having  25,  100,  or  150 


Morse  Relay 


FIG.   9.    Telegraph   Apparatus  Used  at 
ohms  resistance.  Manual  Stations 


TRANSMISSION-MAINTENANCE 

By  John  D.  Taylor 


15.  TRANSMISSION  STANDARDS 

Signal  transmission  is  considered  perfect  in  any  telegraph  circuit  or  connection  if  the 
received  effective  marks  and  spaces  or  dots  and  dashes  are  exactly  the  same  length  as  the 
sent  marks  and  spaces  or  dots  and  dashes.  In  practice,  signals  which  may  be  nearly  per- 
fect as  sent  are  affected  during  transmission  by  circuit  constants,  such  as  the  inductance, 
capacitance,  resistance,  and  leakage  of  the  line  conductors,  by  equipment  characteristics, 
and  by  various  forms  of  interference.  It  has  been  found  that,  with  present  operating 
arrangements,  bias  changes,  large  fortuitous  distortion  (usually  termed  "hits")*  and 
smaller  but  more  frequent  fortuitous  distortion  (see  discussion  in  Article  5)  are  the  princi- 
pal causes  of  transmission  impairment. 

In  designing  telegraph  facilities,  both  line  and  equipment,  the  general  problem  is  to 
provide  a  satisfactory  grade  of  service  in  the  most  economical  and  convenient  manner  that 
will  meet  public  and  operating  needs.  The  trend  is  toward  automatic  transmission  in  the 
telegraph  just  as  it  is  in  the  telephone  field,  in  order  to  increase  the  speed  of  service  most 
efficiently. 

Because  of  the  many  types  of  facilities  employed  in  the  telegraph  plant,  different  tele- 
graph circuits  affect  telegraph  signals  differently,  and,  in  order  for  the  signal  transmission 
to  be  satisfactory  in  any  given  circuit  layout,  it  is  necessary  to  know  in  advance  what 
these  effects  will  be. 

Since,  in  the  present  state  of  the  art,  distance  is  not  a  limiting  factor  in  the  transmission 
of  telegraph  signals,  direct  telegraph  circuits  may  be  provided  from  any  point  to  any  other 
point  in  the  world  by  choosing  the  proper  equipment  and  transmitting  media  for  such 
circuits. 


18-54 


TELEGRAPHY 


16.  TRANSMISSION  COEFFICIENTS 

One  of  the  large  communication  companies  in  the  United  States  in  1926  developed  a 
system  (since  improved)  of  transmission  ratings  of  telegraph  circuits  and  equipment, 
based  on  signal-distortion  measurements  and  on  experience  gained  from  operating  per- 
formance. This  system  is  based  on  the  fact  that,  since  the  distribution  of  distortions 
follows  the  normal  distribution  law,  ratings  or  coefficients,  chosen  as  proportional  to  the 
mean-squared  values  of  the  distortions,  could  be  added  directly  to  give  the  overall  co- 
efficient for  any  combination  of  circuit  units  for  which  coefficients  were  available. 


•4.9- 


orofc 


Doard 
rcuit 


Voice 
frequency 
In  cable        short 
subscriber* 
line 


,  Subscriber" 

station  in 
Baltimore,  Md 


©asper,  Wyo, 


Denver,  CoL  Chicago,  III,  New  York,  N~.Y.  Baltimore^  M<L 

Numerals  are  transmission,  coefficients 

FIG.  1.     Diagram  of  Typical  Teletypewriter  Exchange  Service  Connection  Requiring  a  Regenerative 
Repeater  (Courtesy  Bell  System) 

The  magnitude  of  these  coefficients  was  selected  so  that,  for  satisfactory  signal  trans- 
mission, the  overall  combined  coefficient  should  not  exceed  a  value  of  10.  Where  this 
value  was  exceeded  in  a  given  circuit  layout,  it  would  be' necessary,  by  some  means,  to 
reduce  the  signal  distortion.  For  start-stop  telegraph  equipment,  the  regenerative  re- 
peater is  available  for  correcting  signal  distortion.  It  is  customary  to  insert  them  in  a 
circuit  or  at  the  junction  of  circuits  to  limit  the  overall  coefficient  to  10,  as  shown  in  Fig.  1. 

Duplex  telegraph  repeaters  of  the  differential  type  are  employed  at  intermediate  points 
in  long  d-c  telegraph  circuits  with  ground  return  to  increase  signal  strength.  These  re- 
peaters are  usually  spaced  about  250  miles  apart,  although  this  distance  varies  over  a 
comparatively  wide  range,  depending  on  the  types  of  line  facilities,  equipment,  interfer- 
ence, operating  speeds,  and  other  factors  involved.  These  intermediate  repeaters  do  not 
correct  distortion  but  repeat  the  signals  through  from  section  to  section.  For  this  reason, 
regenerative  repeaters  or  other  distortion-correcting  devices  are  generally  required  every 
two  or  three  repeater  sections. 

17.  CROSSFIRE 


Crossfire  neutralization  between  polar  duplex  grounded  telegraph  circuits  is  necessary 
where  this  form  of  electrical  interference  becomes  objectionable.  As  the  speed  of  opera- 
tion increases,  crossfire  between  such  circuits  assumes  greater  importance. 


Sending  end 


WIrel 


Filter 


Receiving  encf 


-     >Une  current 
—— •*•  Crossfire  current 
*•«•*' Neutrallztpg  current 

FIG.  2.     Neutralization  of  Crossfire  Current  at  the  Sending  Terminal  (Courtesy  Bell  System) 


MAINTENANCE 


18-55 


Crossfire  is  caused  by  the  mutual  inductance  and  capacitance  between  the  telegraph 
circuits  and  by  leakage.  Though  the  major  part  of  this  interference  is  due  to  the  close 
physical  relation  between  the  paralleling  open  wires  and  cable  conductors,  some  of  it  re- 
sults from  couplings  in  equipment  common  to  two  or  more  of  the  paralleling  conductors, 
as  in  composite  sets,  line  filters,  and  loading  coils.  In  general,  the  magnitude  of  the  inter- 
ference is  proportional  to  the  length  of  the  line  wire  and  cable  conductor  parallel.  In  cable, 
the  coupling  is  much  greater  than  in  open  wire,  and,  if  more  than  two  duplex  telegraph 
circuits  are  derived  from  one  quad,  the  crossfire  usually  becomes  prohibitive,  even  for 
comparatively  short  distances,  unless  neutralization  is  applied. 

Figure  2  shows  one  method  of  neutralizing  crossfire  currents  at  the  sending  end  without 
affecting  these  currents  at  the  receiving  end  of  a  polar  duplex  circuit.  Figure  3  shows  one 


Sencflng 


Filter 


Receiving  end 


— >-Llne  current 

— ^-Crossfire  current 

^^Neutralizing  current 

Fia.  3.     Neutralization  of  Crossfire  Current  at  the  Receiving  Terminal  (Courtesy  Bell  System) 

method  of  neutralizing  crossfire  currents  at  the  receiving  end  without  affecting  these  cur- 
rents at  the  sending  end  of  a  polar  duplex  circuit.  In  some  cases,  neutralization  at  the 
sending  or  the  receiving  end  only  will  be  sufficient;  in  the  more  severe  cases,  both  methods 
will  be  required. 

For  sending  end  neutralization,  a  properly  adjusted  condenser  is  employed  between  the 
artificial  lines  of  the  two  circuits  shown,  causing  currents  to  be  set  up  between  them  in  an 
opposing  direction  to  the  crossfire  currents.  For  receiving  end  neutralization,  a  trans- 
former is  inserted  in  the  apexes  of  the  two  duplex  sets,  which,  by  proper  poling  and  adjust- 
ment of  the  coupling,  sets  up  currents  opposing  the  crossfire  currents. 


18.  MAINTENANCE 

The  proper  maintenance  of  telegraph  facilities  plays  a  vital  part  in  providing  the  public 
with  telegraph  service  of  a  satisfactory  grade.  For  this  purpose,  various  types  of  test- 
boards  and  testing  equipment  have  been  developed  and  routines  established  to  insure 
that  the  facilities  are  properly  maintained. 

TESTING  EQUIPMENT  developed  for  maintaining  telegraph  facilities  includes  such 
principal  units  as: 

1.  Monitoring  machines  with  sending  and  receiving  units,  for  checking  the  transmission 
of  telegraph  signals. 

2.  Test  distributors,  sending  substantially  perfect  signals  for  lining  up  and  testing  tele- 
graph circuits  and  apparatus.    One  type  is  for  the  central-office  and  another  for  portable  use. 

3.  Automatic  multiple  senders,  providing  sources  of  battery  reversals  and  of  test  signals 
for  teletypewriter  circuits. 

4.  Telegraph  signal  biasing  sets,  providing  sources  of  biased  battery  reversals  and  tele- 
typewriter test  signals  of  the  inverse  neutral  type;  also  providing  for  measuring  bias  from 
a  distant  sending  end. 

5.  Telegraph  transmission  stability  test  set,  giving  quantitative  indications  on  a  recording 
meter  of  the  freedom  from  bias  of  a  series  of  received  reversals. 

6.  Telegraph  station  test  set,  for  testing  at  outlying  stations  or  small  central  offices,  to 
indicate  distortion  of  received  signals  and  assist  in  determining  wave-shaping  network 
requirements. 

7.  Telegraph  transmission  measuring  set,  indicating  directly  on  meters  the  distortion  in 
signal  reversals  from  start-stop  machines.    One  meter  indicates  bias  (average)  distortion 
and  another  meter  the  peak  value  of  the  total  distortion  (instantaneous  sum  of  bias,  char- 
acteristic, and  fortuitous  effects) .    One  type  of  set  is  for  the  central  office  and  another  is 
for  portable  use. 

8.  Telegraph  crossfire  test  set,  for  determining  the  proper  values  of  capacitance  and  re- 
sistance for  neutralizing  sending-end  crossfire  and  the  proper  inductance  couplings  for 
neutralizing  receiving-end  crossfire  between  grounded  polar  duplex  telegraph  circuits. 


18-56  TELEGRAPHY 

9.  Hit  indicators,  giving  locked  in  indications  of  short-duration  line  disturbances,  suffi- 
ciently large  to  cause  or  almost  cause  the  receiving  relay  to  leave  its  marking  contact,  or 
when  an  initial  spacing  impulse  of  a  message  is  received  from  the  line. 

10.  Qrienttftion-testing'indicator  {portable  type),  for  adjusting  the  orientation  of  a  certain 
type  of  regenerative  repeater, 

11.  Hit  suppressor  unii^  for  preventing  certain  forms  of  interference  on  private-line 
telegraph  facilities  from  being  transmitted  in  a  certain  direction  beyond  the  line  section 
or  sections  in  which  the  hits  occur. 

12.  Carrier  telegraph  test  set,  for  maintaining  carrier  telegraph  systems,  including  fila- 
ment circuit  tests  and  a  drift  measuring  circuit  for  compensator  relay  bias  adjustments. 

13.  Frequency-measuring  devices,  for  checking  carrier  frequencies  used  in  carrier  tele- 
graph systems. 

BIBLIOGRAPHY 

General  References 


1.  Bell  Telephone  System  Practices. 

2.  A.I.E.E.  Transactions. 


Special  References 


1.  F.  E.  D'Humy  and  P.  J.  Howe,  American  Telegraphy  After  100  Years,  AJ.E.E.  Trans.,  Vol.  63, 

1944. 

2.  O.  E.  Pierson,  The  Western  Union  Varioplex  Telegraph  System,  Electrical  Communication,  Vol. 

22,  No.  2. 

3    C.  H.  Cramer,  Some  Modern  Techniques  in  Ocean  Cable  Telegraphy,  AJ.E.E.  Technical  Paper 
47-79,  December  1946. 


RADIO  TELEGRAPH  SYSTEMS 

By  J.  L.  Finch 

In  radio  telegraph  communications  it  is  necessary  to  have  a  transmitter,  and  a  receiver 
and  an  operator's  position.  For  the  less  important  circuits  these  are  often  located  together 
under  the  control  of  one  operator  who  alternately  sends  and  receives  messages.  This 
method  of  operation  is  known  as  the  simplex  method.  For  more  important  circuits  the 
transmitter  and  the  receiver  are  arranged  for  simultaneous  operation,  and  this  is  known 
as  the  duplex  method.  In  the  duplex  method  it  is  usually  advisable  to  have  the  trans- 
mitter and  receiver  located  at  different  points  to  reduce  interference  troubles.  This 
arrangement  becomes  increasingly  important  as  more  transmitters  and  receivers  are  used. 
When  it  is  necessary  to  locate  the  transmitters  and  receivers  at  the  same  point,  as  on  ship- 
board, special  precautions  must  be  taken  to  prevent  mutual  interference  and  the  frequency 
separation  between  transmitted  and  received  signals  must  be  kept  relatively  great. 

When  ratio  communication  is  carried  on.  to  and  from  large  cities  it  is  the  usual  practice 
to  have  a  central  office  in  the  city  with  both  the  transmitting  and  the  receiving  centers 
located  well  outside  the  city  and  connected  by  wire  lines  or  by  ultra-high-frequency  radio 
control  circuits. 

19.  CHOICE  OF  TRANSMITTER  AND  RECEIVER  SITES 

The  sites  are  usually  chosen  where  ample  level  space  is  available  for  directional  antennas 
for  each  communication  circuit  planned.  The  availability  of  a  dependable  power  supply 
and  of  reliable  control  channels  is  a  further  consideration.  The  receiving  centers  are 
usually  located  where  man-made  static  is  low.  It  is  desirable  that  hills  and  mountains  in 
front  of  the  antennas  do  not  rise  at  an  angle  of  more  than  3°  for  long-distance  communica- 
tion at  short  waves,  although  angles  of  as  much  as  5°  can  usually  be  tolerated.  For 
shorter  distances  the  angles  may  be  greater.  The  land  immediately  in  front  of  an  antenna 
has  a  direct  bearing  on  the  vertical  directivity  and  should  be  level.  For  long  waves,  moun- 
tains and  valleys  within  a  fraction  of  a  wavelength  have  little  effect  on  the  antenna  direc- 
tivity. 

20.  CHOICE  OF  FREQUENCIES 

The  most  desirable  frequency  to  be  used  between  any  two  points  depends  upon  many 
factors  which  vary  with  time  of  day,  season  of  year,  position  in  the  sunspot  cycle,  occur- 
rence of  magnetic  storms,  and  location  of  the  great-circle  path  over  which  the  signals  must 


REDUCTION  OF  FADING  EFFECTS  18-57 

travel  with  respect  to  the  aurora  areas  centered  about  the  earth's  magnetic  poles.  These 
factors  are  mostly  related  to  wave  propagation  and  are  described  in  article  10-23.  Usable 
frequencies  start  at  about  10  kc  and  extend  as  high  as  30,000  Me.  The  very  low  frequen- 
cies are  propagated  effectively  and  are  relatively  steady  and  consistent,  but  they  require 
very  large  and  expensive  antenna  systems,  particularly  at  the  transmitting  end,  in  order 
to  radiate  even  a  small  percentage  of  the  generated  power.  At  the  receiving  end  extensive 
antennas  are  required  to  get  good  directivity  and  thus  a  favorable  signal-to-noise  ratio. 
Further,  the  transmitting  antennas  must  be  very  sharply  tuned  in  order  to  be  efficient,  and 
this  fact  limits  their  modulation  capabilities  to  relatively  low  keying  rates.  It  should  be 
noted  that  propagation  at  frequencies  from  50  kc  to  30  Me  is  subject  to  wide  fluctuation. 

Commercial  companies  have  found  it  worth  while  to  maintain  and  operate  long-wave 
facilities  which  are  already  in  existence  for  use  during  magnetic  storms  and  to  serve  for 
re-establishing  contact  with  the  remote  points  after  short  waves  have  suddenly  faded  out. 

Frequencies  between  3  and  25  Me  have  been  found  the  most  useful  for  long-distance 
communications.  Those  between  3  and  10-13  Me  are  useful  over  these  long  distances 
when  most  or  all  of  the  radio  path  is  in  darkness  while  those  from  10-13  to  25  Me  are 
useful  when  most  or  all  of  the  path  is  in  daylight.  The  lower  of  these  frequencies  are 
useful  over  shorter  distances  in  the  daytime.  The  same  frequencies  can  be  used  succes- 
sively in  different  parts  of  the  world  as  the  daylight  and  darkness  areas  progress  around 
the  earth. 

Frequencies  above  30  Me  are  rarely  useful  for  extensive  periods  for  long  distances  (over 
2500  miles)  and  those  above  50  Me  can  be  relied  upon  as  a  rule  not  to  carry  to  distant 
points  on  the  earth's  surface  and  so  can  be  used  over  and  over  again  on  circuits  separated 
by  only  a  few  hundred  miles.  Because  signals  at  these  higher  frequencies  will  not  bend 
very  much,  an  optical  path  or  one  approaching  an  optical  path  between  the  transmitting 
and  receiving  antennas  is  necessary. 

21.  REDUCTION  OF  FADING  EFFECTS 

The  use  of  short  waves  for  communicating  over  long  distances  is  very  frequently  beset 
with  fading  caused  by  the  alternate  addition  and  subtraction  of  the  signal  arriving  over 
different  ether  paths.  This  is  due  to  the  varying  phase  relation  of  the  arriving  radio- 
frequency  waves.  It  has  been  found  that  much  of  this  fading  is  very  local  in  character. 
When  a  signal  has  faded  out  at  one  location  it  may  be  coming  in  at  full  strength  a  few 
hundred  feet  away.  Also  when  at  a  given  point  one  radio  frequency  has  faded  out  com- 
pletely, another  frequency  only  a  few  hundred  cycles  different  being  radiated  from  the 
same  transmitting  antenna  may  be  corning  in  at  full  strength  at  this  point.  Receiving 
systems  which  employ  two  or  more  receiving  antennas  spaced  apart  geographically  are 
known  as  space  diversity  receiving  systems.  The  telegraphic  signals  received  by  each  antenna 
are  rectified  and  passed  through  a  limiter  which  cuts  off  the  tops  of  all  the  characters  that 
would  exceed  a  certain  value.  The  various  signals  are  then  combined  in  such  a  manner 
that  all  of  them  must  fade  at  once  to  cause  a  signal  failure.  When  using  three  separated 
receiving  antennas  this  diversity  effect  reduces  the  failures  due  to  fading  to  a  very  small 
fraction  of  that  suffered  by  the  signal  received  on  any  one  of  the  antennas.  Systems  that 
transmit  two  or  more  slightly  different  radio  frequencies  from  one  antenna  are  termed 
frequency  diversity  systems.  In  this  case  the  same  receiving  equipment  is  used  as  otherwise, 
since  the  different  radio  frequencies  being  received  are  so  close  together  that  the  receiver 
responds  to  all  of  them  substantially  equally.  The  limiting  used  is  such  that  all  frequencies 
must  fade  out  at  once  to  cause  a  signal  failure.  This  system  results  in  a  very  marked  in- 
crease in  the  reliability  of  the  circuit. 

To  obtain  the  different  frequency  components  required  in  the  transmitter  output  for 
frequency  diversity  it  is  usual  to  phase-modulate  the  carrier  at  an  audio  rate  of  about 
600  cycles  and  with  a  maximum  phase  deviation  of  about  1  radian.  This  modulation 
results  in  the  production  of  two  side  frequencies  spaced  600  cycles  each  side  of  the  carrier. 
The  amplitude  of  each  side  frequency  is  a  little  less  than  half  the  amplitude  of  the  unmodu- 
lated carrier.  Phase  modulation  can  be  accomplished  quite  simply,  and  it  does  not  neces- 
sitate reduction  of  the  total  power  generated  by  the  transmitter. 

It  is  common  practice  to  key  two  different  transmitters  operating  at  different  frequen- 
cies with  the  same  signals  and  to  combine  the  outputs  of  two  receivers  at  the  traffic  office 
to  insure  against  service  interruptions,  particularly  at  the  time  of  day  when  one  of  the 
frequencies  is  about  to  fade  out  and  it  is  desired  to  replace  it  with  another. 


18-58  TELEGRAPHY 


22.  RADIO  INTERFERENCE 

Interference  between  radio  channels  should  be  eliminated  by  spacing  the  frequencies 
sufficiently  far  apart  and  by  making  the  receivers  selective  enough  to  differentiate  between 
the  desired  and  undesired  signals.  The  radiation  at  unauthorized  frequencies  such  as  at 
harmonics  of  the  desired  frequency  wave  must  be  avoided.  In  some  types  of  transmit- 
ters, frequencies  lower  than  the  desired  ones  are  generated  and  then  the  desired  harmonic 
is  radiated.  The  radiation  of  these  lower  frequencies  must  be  avoided.  Vacuum-tube 
transmitters  are  prone  to  generate  spurious  or  parasitic  frequencies.  The  radiation  of 
such  frequencies  must  be  avoided.  These  parasitic  frequencies  often  modulate  the  desired 
frequency  wave,  resulting  in  modulation  sidebands  which  will  be  radiated  with  the  desired 
wave  if  allowed  to  exist  and  will  cause  interference  on  other  channels.  Often  when  a 
transmitter  is  free  from  parasitic  oscillations  in  the  steady  state,  both  marking  and  spac- 
ing, such  oscillations  occur  during  the  transient  period  at  the  beginning  and  end  of  signal- 
ing characters.  These  oscillations  are  known  as  key-click  parasitic  oscillations.  Other 
interchannel  interference  sometimes  results  from  the  modulation  products  or  side  fre- 
quencies generated  by  the  keying,  these  being  particularly  noticeable  at  high  signaling 
speeds  and  when  the  characters  are  square  ended.  To  round  the  characters  to  reduce  these 
side  frequencies  when  using  efficient  class  C  amplifiers  in  the  transmitter  and  with  on-off 
keying  is  difficult  and  expensive.  When  using  "frequency  shift"  or  "two-tone"  keying, 
however,  the  signal  characters  can  be  rounded  quite  simply  and  the  objectionable  side- 
frequency  radiations  greatly  reduced.  Two  transmitters  at  the  same  station,  particularly 
when  operating  at  closely  spaced  frequencies,  sometimes  intermodulate  each  other  and 
radiate  signals  in  adjacent  bands.  This  can  usually  be  made  negligible  by  reducing  the 
cross  coupling  between  the  two  systems. 

Receivers  may  cause  interference  with  other  receivers,  usually  due  to  radiation  from  the 
first  heterodyne  oscillator  into  the  room  and  into  the  receiver  power  wires  and  into  the 
receiving  antenna.  It  is  particularly  important  to  avoid  radiation  back  into  the  antenna 
of  a  level  of  more  than  1  microvolt  when  one  antenna  is  used  with  a  number  of  receivers 
at  various  and  variable  frequencies. 

23.  FREQUENCY  SHIFT  BUYING 

It  has  been  the  general  practice  in  the  past  to  key  transmitters  by  interrupting  the 
transmitter  output  power.  Recently  equipment  has  been  developed  to  take  advantage  of 
the  "frequency  shift  keying"  system,  also  known  as  the  "two-tone  keying"  system.  This 
method  employs  one  radio  frequency  for  "mark"  and  a  second  radio  frequency  for  "space," 
each  at  the  same  power.  The  separation  between  the  two  is  not  critical.  At  present  a 
separation  of  850  cycles  is  in  common  use.  To  reduce  key-click  interference  in  adjacent 
channels  it  is  desirable  to  shift  the  frequency  of  a  single  oscillator  and  at  a  rate  only  suffi- 
cient to  accommodate  the  required  keying  speed. 

In  receiving  frequency  shift  signals,  amplitude  variations  are  limited  out  by  the  use  of 
a  limiter  stage.  The  resulting  signals  of  fixed  amplitude  and  variable  frequency  are  im- 
pressed upon  a  discriminator  the  output  of  which  may  operate  a  relay  device,  tone  keyer, 
or  other  utilization  device.  The  limiter  stage  largely  eliminates  the  effect  of  atmospheric 
or  other  interference  whose  strength  is  lower  than  3  db  below  the  signal. 

Results  have  indicated  that  in  practice  the  transmitter  power  can  be  reduced  as  much 
as  10  db,  after  adopting  the  frequency  shift  keying  system,  without  degrading  the  service. 

24.  TRAFFIC  OFFICE  AND  EQUIPMENT 

In  central  offices  carrying  small  volumes  of  traffic,  particularly  over  radio  circuits  of 
inferior  reliability,  it  is  advantageous  to  have  the  transmitting  and  receiving  positions 
located  adjacent  to  each  other.  This  is  for  convenience  in  asking  and  giving  acknowledg- 
ments and  for  "breaking"  the  circuit.  For  handling  larger  volumes  of  traffic  over  a  num- 
ber of  reliable  circuits,  it  is  advantageous  to  arrange  the  transmitting  positions  in  one 
group  and  the  receiving  positions  in  another. 

Intelligence  can  be  transmitted  over  radio  telegraph  circuits  in  the  Morse  code  or  by 
means  of  automatic  printers  using  their  own  special  codes.  For  the  simplest  equipment 
Morse  signals  are  transmitted  by  hand  and  received  aurally  and  transcribed  by  hand  or 
on  a  typewriter.  For  higher  speeds  and  more  reliable  service  automatic  sending  and  re- 
ceiving is  used.  The  messages  are  punched  on  a  typewriter-like  machine  known  as  a 


TEAFFIC   OFFICE  AND  EQUIPMENT  18-59 

perforator.  A  sample  tape  is  shown  in  Fig.  1.  The  tape  is  run  through  an  auto-head  device 
which  has  electrical  contacts  and  which  form  the  Morse  characters  corresponding  to  the 
letters  punched.  The  received  signals  operate  an  ink  tape  recorder.  These  recorders  com- 
monly involve  a  moving  tape  with  a  pen  pressed  against  it.  The  pen  is  retained  in  its 
lower  position  for  "space"  and  in  its  upper  position  for  "mark,"  making  a  record  as  shown 
in  Fig.  1.  Automatic  Morse  communication  is  practical  at  speeds  of  20  to  500  words  per 
minute. 

For  average  speeds  up  to  40  or  50  words  per  minute  one  operator  at  the  transmitting 
end  perforates  the  tape  and  tends  to  the  auto-head  machine.  At  the  receiving  end  one 
operator  views  this  tape  as  the  message  is  recorded  and  transcribes  the  messages  on  a 
typewriter.  For  higher  speeds  of  transmission  the  tape  must  be  punched  by  two  or  more 
operators  and  sections  fed  successively  into  the  auto-head.  The  recorder  tape  must  be 
divided  between  two  or  more  receiving  operators  for  transcription. 

Automatic  printers  are  advantageous  in  that  they  make  it  unnecessary  for  the  operators 
to  learn  the  code,  they  reduce  human  errors  in  transcribing  from  code  signals,  and  they 
permit  automatic  transcription  of  incoming  messages  directly  in  a  form  suitable  for  de- 
livery to  the  customer.  Their  use,  however,  eliminates  the  possibility  of  detecting  errors 

Auforrraflc  Transmitter  Tape  __^__ 


nn  noon  nnnn  nnn  n  nnnn  nnn 


Recorder  Tape 
FIG.  1.     Tapes  for  Automatic  Sending  and  Receiving 

due  to  mutilation  of  the  signals  introduced  in  the  radio  path.  Frequently  a  trained  opera- 
tor can  transcribe  correctly  a  mutilated  signal  such  as  would  cause  a  printer  to  make  an 
error  or  to  operate  an  error-indicating  device.  Further,  the  transmission  speed  is  limited 
to  that  at  which  a  printer  will  function  properly,  i.e.,  to  speeds  of  60  to  100  words  per 
minute.  The  printers  in  general  use  employ  a  five-unit  code  and  respond  to  any  combina- 
tion of  signals  whether  mutilated  or  not.  An  error-indicating  printer  is  in  use  which 
employs  a  seven-unit  code.  All  operations  use  three  marking  and  four  spacing  elements. 
Any  other  combination  will  operate  an  error-indicating  device. 

In  connection  with  radio  telegraphy  it  is  customary  to  rate  transmission  speeds  in  words 
per  minute,  each  word  consisting  of  an  average  of  five  letters.  More  accurately,  each  word 
consists  of  48  units  *  or  minimum  elements,  each  unit  being  taken  as  the  length  of  one 
dot  or  one  space  between  dots,  and  the  length  of  dashes  and  spaces  between  letters  being 
equal  to  three  units.  From  this  it  can  be  calculated  that  when  the  auto-head  is  running 
at  a  rate  to  make  40  dots  per  second  the  transmission  rate  will  be  100  words  per  minute. 
Similarly  printer  speeds  are  calculated  on  the  basis  of  six  operations  per  word,  allowing 
five  characters  and  one  space  per  word. 

MULTIPLEX.  In  order  to  make  use  of  the  high  signaling  speed  capabilities  of  short- 
wave radio  circuits  and  still  enjoy  the  advantage  of  automatic  printer  operation  at  nominal 
speeds,  means  have  been  devised  for  dividing  the  total  time  available  for  transmission  be- 
tween two  or  more  channels.  This  system  is  known  as  time  division  multiplex  (see  article  7) . 
This  system  can  also  be  used  for  Morse  operation;  it  is  advantageous  because  each  channel 
can  be  copied  as  it  comes  in,  and  thus  delays  in  message  delivery  caused  by  the  necessity 
of  dividing  tape  received  at  high  speeds  between  two  or  more  operators  can  be  avoided. 

MULTICHANNEL.  A  second  method  of  obtaining  more  than  one  telegraph  channel 
on  a  single  radio  circuit  utilizes  single-sideband  equipment  such  as  has  been  developed 
primarily  for  telephone  service.  Such  a  system  can  carry  a  large  number  of  individual 
tones,  each  keyed  as  desired  with  printer  signals  or  Morse  signals  and  each  separated  from 
the  others  at  the  receiving  end  by  means  of  wave  filters. 

Both  the  multiplex  and  the  multichannel  systems  are  more  expensive  initially  and  more 
costly  to  maintain  and  operate  than  single-channel  systems,  but  they  are  worth  while 
when  large  volumes  of  traffic  are  to  be  handled  because  they  save  space  in  the  radio  spec- 
trum and  at  the  same  time  provide  for  printer  operation  or  Morse  operation  at  speeds  that 
can  be  transcribed  currently. 

*  The  term  "baud"  has,  in  some  sections  of  the  industry,  been  applied  to  these  units.  In  the  offi- 
cially correct  usage  the  baud  is  the  unit  of  telegraphic  speed  or  of  rapidity  of  modulation  corresponding 
to  one  minimum  element  per  second. 


18-60 


TELEGKAPHY 


25.  CONTROL  CHANNELS 

The  signals  must  be  carried  from  the  central  office  to  the  transmitting  station,  and  from 
the  receiving  station  to  the  central  office,  over  suitable  control  channels.  These  channels 
can  be  carried  over  wire  lines  or  over  ultra-high-frequency  radio  circuits.  When  wire  lines 
are  available  d-c  signals  may  be  used,  either  unidirectional  or  polarized  (plus  and  minus), 
or  keyed  tone  signals  may  be  employed.  The  tone  signals  have  the  advantage  that  a 
number  of  them  can  be  carried  over  one  tone  pair  and  separated  by  wave  niters  (see  art- 
icle 11-10).  When  ultra-high-frequency  radio  control  circuits  are  used,  keyed  tone  signals 
are  employed.  A  number  of  similar  groups  of  tone  signals  can  be  transmitted  over  a  single 
radio  circuit  by  a  system  similar  to  the  voice-carrier  system  as  described  in  article  17-9. 

TONE  KEYERS.  The  keying  device  or  tape  transmitter  device  in  the  central  office 
usually  closes  an  electric  circuit  by  means  of  a  pair  of  contacts  for  marking  and  opens  it 
for  spacing.  This  action  is  made  to  key  a  tone  by  means  of  a  tone  keyer  such  as  that 
shown  in  Fig.  2.  The  equivalent  results  can  be  accomplished  by  purely  electronic  means 


kD-c  transient 
'      balancex 


Key 


-i20V  ^    +380  V 

PIG.  2.     Circuit  Diagram  of  a  Tone  Keyer 

applying  the  same  general  principles.  Tone  keyers  are  also  used  at  the  receiving  station 
by  means  of  which  the  incoming  signal  is  made  to  key  a  tone  for  transmission  to  the  central 
office. 

TONE  SIGNAL  CONVERTERS.  At  the  transmitting  station  the  tone  signals  must 
be  converted  into  d-c  signals  to  key  the  transmitter.  The  device  for  accomplishing  this  is 
known  as  a  tone  signal  converter  and  may  take  the  form  shown  in  Fig.  3.  This  embodies  a 


-4 .V -i--, 


J _  _-400V         \ 

Tone  slgnaj  convorter  Threshold 

FIG.  3.     Circuit  Diagram  of  a  Tone  Signal  Converter 

threshold  device  for  suppressing  low-level  noise  and  for  preventing  tails  on  the  characters 
from  causing  "heavy"  signals.  It  embodies  a  Umiter  for  cutting  off  the  tops  of  the  signals 
and  a  low-pass  filter  to  smooth  out  the  rectified  tone  and  thus  to  keep  the  signals  from 
being  modulated  by  the  tone  itself.  The  gain  control  is  used  for  setting  the  level  of  the 


CONTKOL  CHANNELS 


18-61 


tone  signals,  which  usually  have  been  rounded  by  passing  through  the  wave  niters  of  the 
tone  channel  to  give  the  correct  "weight"  of  signals  in  the  transmitter.  Lowering  the 
gain  makes  the  signals  lighter,  and  raising  it  makes  them  heavier. 

RECORDER  DRIVE.  At  the  central  office  the  incoming  tone  signals  must  be  con- 
verted to  direct  current  to  drive  the  ink  tape  recorder  or  printer  relay.  The  device  for 
accomplishing  this  may  take  the  form  shown  in  Fig.  4.  A  simple  tone  rectifier  circuit  will 
usually  suffice  for  driving  the  printer  relay  for  printer  service. 


Rfectlfier 


Recorder 


. ^  ^  :fc350V 

'"Space"  currenis          '"Made"'  currents 

FIG.  4.     Circuit  Diagram  of  a  Recorder  Driving  Unit 

BANDWIDTHS.  The  transmission  of  keyed  tone  signals  requires  a  definite  band 
which  must  have  a  width  roughly  proportioned  to  the  keying  speed.  Satisfactory  service 
may  be  achieved  by  allowing  for  the  transmission  of  no  more  than  the  first  side  frequency 
of  the  keying  speed  under  stable  input  conditions  and  stable  transmission  characteristics 
such  as  are  normally  achieved  in  first-class  lines.  Thus  for  125  words  per  minute  Morse, 
i.e.,  a  keying  frequency  of  50  dots  per  second,  the  usable  channel  width  must  be  at  least 
100  cycles.  For  unstable  conditions  it  is  desirable  to  transmit  the  first,  second,  and  third 
side  frequencies.  Thus,  a  100-cycle  band  width  will  be  suitable  for  speeds  of  42  words 
per  minute.  In  actual  practice  100-cycle  bands  have  been  found  suitable  for  speeds  up  to 
60  words  per  minute. 


SECTION  19 
FACSIMILE  TRANSMISSION  AND  RECEPTION 


BY 

MAURICE  ARTZT 


ART.  SCANNING  SYSTEMS  PAGE 

1.  Picture  Elements 02 

2.  Scanners 03 

3.  Scanner  Amplifiers 07 

RECORDING  SYSTEMS 

4.  Photographic  Recording 11 

5.  Wet  Electrolytic  Recording 12 

6.  Dry  Electrolytic  Recording 13 

7.  Carbon-paper  Recording IS 

8.  Comparison  of  Recording  Methods 16 

9.  Recording  Amplifiers 16 


j^^  SYNCHRONIZING  AND  PHASING  PAGB 

10.  Synchronizing 18 

11.  Phasing 21 

TRANSMISSION  CHARACTERISTICS 

12.  Wire  Line  Transmission 22 

13.  Radio  Transmission 22 

SPECIALIZED  APPLICATIONS 

14.  Duplicators 23 

15.  Tape  Facsimile 23 


19-01 


FACSIMILE  TRANSMISSION  AND  RECEPTION 

By  Maurice  Artzt 

SCANNING  SYSTEMS 


Facsimile  is  defined  to  include  all  systems  whereby  a  picture  is  broken  into  separate 
picture  elements,  these  elements  being  transmitted  by  some  connecting  means  to  a  distant 
recorder  where  they  are  reassembled  into  their  original  positions  to  form  a  copy  of  the 
original.  The  word  "picture"  in  the  above  statement  includes  also  diagrams,  typing, 
handwriting,  photographs,  and  any  other  form  of  printed  or  written  material. 

Three  distinct  operations  are  performed  in  the  transmitting  and  recording  of  facsimiles: 
first,  the  breaking  up  of  the  picture  in  some  orderly  manner  into  its  separate  elements  of 
varying  shades,  this  process  being  called  scanning;  second,  the  transmitting  of  these  ele- 
ments to  the  recorder  by  means  of  signals  arranged  to  represent  the  electrical  equivalent  of 
these  elements;  third,  the  rebuilding  of  these  signals  by  a  recorder  into  a  printed  copy  of  the 
original  by  a  reversal  of  the  scanning  process. 

A  fourth  part  of  a  facsimile  system,  supplementary  but  very  necessary,  is  a  method  of 
synchronizing  the  recorder  and  scanner.  The  timing  of  the  signals  received  must  agree 
exactly  with  the  timing  of  the  recorder,  in  phase  as  well  as  frequency,  if  the  copy  received  is 
to  be  undistorted. 

In  the  following  articles  the  terms  used  are  in  accordance  with  the  definitions  and  stand- 
ards as  set  up  in  1942  by  the  Institute  of  Radio  Engineers.  See  the  first  reference  in  the 
Bibliography. 

1.  PICTURE  ELEMENTS 

In  processing  a  picture  by  facsimile,  the  picture  is  resolved  into  dots,  or  picture  elements, 
similar  to  the  small  dots  used  in  printing  a  picture  in  a  newspaper  or  magazine.  These  dots 
are  obtained  by  "screening"  in  the  printing  process;  they  are  obtained  by  scanning  in 
facsimile. 

Halftones  in  newspaper  work  have  from  60  to  120  dots,  or  picture  elements,  per  inch, 
whereas  fine  magazine  printing  may  use  as  many  as  250  dots  per  inch.  In  facsimile  the 
limits  are  of  about  the  same  order,  almost  all  present  facsimile  systems  using  100  dots  (or 
lines)  per  inch,  as  an  average.  Each  picture  element  in  a  facsimile  is  sent  as  a  separate 
signal.  If  the  number  of  dots  is  too  high,  the  speed  of  transmission  is  very  slow;  if  too  few 
elements  are  used,  the  detail  will  not  be  good  enough.  To  send  a  picture  of  100  dots  per 
inch  requires  as  many  as  10,000  separate  signals  per  square  inch  of  surface  covered. 

Figure  1  illustrates  the  difference  to  be  expected  between  50  and  100  lines  per  inch,  when 
the  subject  matter  is  ordinary  typing.  Some  of  the  type  would  be  unreadable  if  only  50 


(a)  50  lines  per  inch  (&)  100  lines  per  inch. 

FIG.  1.    Difference  in  Detail  for  50  and  100  Lines  per  Inch  for  Typewritten  Letters 

lines  per  inch  were  used,  as  can  be  seen  by  the  poor  formation  of  the  a.  The  100-line-per- 
inch  detail,  though  not  forming  perfect  letters,  leaves  no  doubt  as  to  their  identity.  In 
commercial  facsimile,  letters  from  ordinary  typewriters  often  comprise  the  original,  and 
approximately  100  lines  per  inch  are  necessary  to  insure  readability. 

Dots  per  inch  and  lines  per  inch  are  used  interchangeably  in  the  above  paragraph,  a 
practice  which  is  not  always  permissible.    Some  facsimile  systems  break  the  subject  up  into 

19-02 


SCANNERS 


19-03 


dots  and  send  a  separate  signal  for  each  dot,  whether  white,  black,  or  gray.  Others,  how- 
ever, break  the  sheet  up  into  parallel  lines  and  send  signals  only  for  the  black  areas  as  en- 
countered. Each  line  is  then  a  continuous  signal,  varying  in  intensity  with  the  shading  of 
the  original,  and  not  made  up  of  an  exact  number  of  picture  elements  as  the  dotted  picture 
is.  The  detail  limits  are  the  same  in  either  case,  and  the  maximum  number  of  picture 
elements  per  square  inch  is  the  same. 

These  picture  elements,  as  observed  by  the  scanner  in  the  process  of  transmission,  will  be 
of  two  general  types,  either  of  the  simple  black-and-white  variety,  such  as  typing,  line 
drawings,  and  so  forth,  or  of  the  halftone  variety,  in  which  all  shades  of  gray  from  white 
to  black  may  occur.  Two  separate  types  of  scanners  are  not  necessary,  but  the  amplifier 
equipment  will  sometimes  be  different.  Any  system  capable  of  transmitting  and  recording 
halftones  will  also  operate  properly  on  a  purely  black-and-white  original,  but  the  reverse  is 
not  necessarily  true. 

2.  SCANNERS 

A  facsimile  scanning  system  includes  an  optical-mechanical  scfl.Tm.er  designed  to  project 
a  small  spot  of  light  on  the  subject  copy,  to  gather  the  reflected  or  transmitted  light  from 


-Driving  motor 


Guide  rod 


Drum 


Phototube 


Subject 
copy 


Drum  shaft 


•  Lead  screw  gearing 
FIG.  2.    Scanner  with  Traversing  Optical  System 

the  subject  into  a  phototube,  and  to  bring  all  parts  of  the  subject  under  this  scanning  spot 
in  some  orderly  manner.  The  signals  generated  in  the  phototube  by  the  varying  light  values 
reflected  from  the  copy  are  then  either  amplified  directly  or  processed  in  other  ways  to  form 
a  usable  electrical  signal  of  a  type  suited  for  the  particular  application. 

SCANNING  METHODS.  In  the  simplest  form  of  scanning,  regular  lines  are  "ruled" 
across  the  sheet  by  this  spot  of  light  at  some  particular  number  of  lines  per  inch,  and  signals 
are  sent  out  representing  each  small  area  as  it  is  encountered.  The  sheet  is  thus  broken 


19-04         FACSIMILE   TRANSMISSION  AND  RECEPTION 


into  a  number  of  narrow  lines,  all  of  the  same  width,  and  these  lines  are  transmitted  one 
after  another  until  the  entire  subject  has  been  covered. 

Scanning  is  generally  done  in  only  one  direction  and  seldom  back  and  forth.  There  are 
two  reasons  for  this :  first  a  unidirectional  scanner  is  simpler  to  construct  and  requires  less 
precision  in  gearing,  and  second  the  synchronizing  system  for  back-and-forth  scanning  must 
be  far  more  accurate. 

The  simplest  form  of  the  scanner,  and  therefore  the  one  most  generally  used,  consists  of  a 
drum  upon  which  the  original  subject  matter  is  wrapped,  and  an  optical  system  arranged  to 
project  a  small  spot  of  light  on  the  surface  of  the  paper.  This  spot  is  usually  somewhat 
smaller  than  the  width  of  one  scanning  line.  As  the  drum  is  revolved,  the  optical  system  is 
moved  relative  to  the  drum  the  width  of  one  scanning  line  for  each  revolution  of  the  drum. 
The  entire  subject  is  thus  gradually  passed  under  the  scanning  spot.  See  Fig.  2. 

A  phototube  is  arranged  to  pick  up  the  light  reflected  from  the  surface  of  the  paper,  and 
this  light  reaching  the  phototube  will  be  varied  in  intensity  by  the  different  areas  of  black, 
gray,  and  white  that  may  be  presented  to  view.  The  output  of  the  phototube  will  be  a 
minimum  for  black  and  maximum  for  white  and  will  represent  electrically  the  scanning  of 
the  copy.  This  phototube  output  is  then  applied  to  the  input  of  the  amplifier  system. 

All  motions  in  the  scanning  process  pictured  in  Fig.  2  are  relative.  Thus  the  optical  sys- 
tem may  be  rotated  in  place  of  the  drum,  and  the  motion  along  the  axis  may  be  made  by 
moving  either  the  drum  or  the  optical  system  relative  to  each  other.  All  methods  of 
bringing  about  this  relative  motion  have  been  used. 

In  one  commercial  type  of  scanner  used  for  news  picture  transmission  the  drum  is  re- 
volved and  moved  along  the  axis  by  a  lead  screw  cut  on  its  shaft,  and  the  optical  system  is 
stationary.  See  Fig.  3. 


Motor 


Drum  shaft 
half  lead  screw  and  half  keyway 


Nut  on  lead  screw  moves 
rotating  drum  along  shaft 


Scanning 
lenses 


Drum  with  subject  copy 
FIG.  3.    Scanner  with  Stationary  Optics  and  Drum  Feeding  for  Line  Advance 

As  facsimile  speeds  have  increased,  the  time  required  to  load  the  subject  on  a  drum  has 
become  an  increasingly  greater  proportion  of  the  total  time  of  transmission  of  the  copy. 
Various  ingenious  methods  of  loading  and  scanning  have  been  devised  to  minimize  this 
loss  of  time.  In  one  form  of  scanner  designed  for  rapid  loading,  the  subject  is  wrapped  face 
in  around  a  transparent  cylinder,  and  the  optical  system  is  rotated  inside  this  cylinder. 
See  Fig.  4,  By  this  method  the  scanning  process  does  not  have  to  be  stopped  to  remove  one 
subject  and  put  another  in  place.  Thus  the  time  to  resynchronize  and  rephase  for  the  next 
subject  is  not  lost,  and  the  time  between  succeeding  subjects  is  reduced. 

In  all  scanners  illustrated  thus  far  the  original  must  be  of  such  size  that  it  can  be  properly 
chimped  on  the  scanning  drum.  Thus  copy  width  must  be  approximately  the  circumference 
of  the  drum,  minus  the  separation  between  clamps.  The  length  is  not  so  restricted  and  may 
be  anything  up  to  the  length  of  the  drum.  Another  type  of  scanner  in  which  width  is  not 
restricted  except  as  to  a  maximum  value  is  shown  in  Fig.  5.  The  subject  is  placed  face  in 
on  a  stationary  transparent  semicylinder  and  held  there  by  a  curtain  (not  shown) .  Two 
microscopes  and  light  pick-up  systems  are  rotated  and  traversed  inside  to  scan  the  copy. 
The  two  optical  systems  are  set  exactly  180°  apart  and  in  the  same  plane  so  that  the  signals 


SCANNERS 


19-05 


Ssl 


19-06         FACSIMILE  TRANSMISSION  AND   RECEPTION 

generated  in  the  single  phototube  will  be  the  equivalent  of  that  from  a  single  optical  system 
scanning  a  complete  cylinder.  With  this  scanner,  unloading  and  reloading  takes  only  a  few 
seconds  as  the  copy  is  not  clamped  to  hold  it  in  place.  As  two  optical  systems  are  used  the 
shaft  speed  will  be  one-half  that  of  an  equivalent  drum  scanner. 


In  one  other  type  of  scanner  used  for  telegraph  service  reloading  with  the  next  message  is 
accomplished,  by  dropping  out  the  drum  and  copy  when  scanning  is  completed  and  re- 
loading with  another  drum  containing  the  next  message  from  a  hopper  feed.  This  is  done 
automatically,  and  provision  is  made  for  accommodating  a  number  of  additional  message- 
carrying  drums  so  that  the  messages  follow  one  another  in  rapid  succession. 


SCANNER  AMPLIFIERS 


19-07 


3.  SCANNER  AMPLIFIERS 

The  signals  generated  by  the  phototube  will  vary  in  amplitude  with  the  shading  on  the 
subject  being  scanned,  and  in  frequency  with  the  speed  of  the  scanning  spot  and  the  kind  of 
subject  copy.  The  highest  frequency  will  be  determined  by  the  size  of  the  smallest  "dot" 
it  is  expected  to  transmit;  the  lowest  frequency  will  be  zero  or  a  direct  current  to  represent 
the  large  areas  of  white  or  black  encountered  in  nearly  all  types  of  copy. 

FREQUENCY  SPECTRUM.  The  highest  frequency  is  determined  as  follows:  Take 
the  width  of  the  smallest  "dot"  it  is  expected  to  transmit  and  rule  a  pattern  of  lines  of  this 
width,  separating  them  by  the  width  of  the  dot.  If  the  scanning  is  to  be  at  100  lines  per  inch 
then  these  lines  will  be  0.01  in.  in  width,  50  to  the  inch,  and  separated  by  0.01  in.  Such  a 
pattern  is  shown  in  Fig.  6 A.  The  fundamental  keying  frequency  of  the  phototube  current 


SCANNING 
LINE 


A.      SCANNING   PATTERN    FOR    FUNDAMENTAL   KEYING 

FREQUENCY   GIVING   GREATEST   APERTURE    DISTORTION 


PHOTOTUBE 
SIGNAL 


OTUBE    i     Tl          I         I    _    I         I          I 

«"  ILLmJ 


f>    .   E    ,  2E  Ps in  fiT     Sin  3mT  ,  Sl'n  5g/T         _     J 
*    -   2+Tf    L     I        "       3         +        5  ] 

B.      PERFECT  SIGNAL  WITH  APERTURE  OF  INFINITESIMAL  WIDTH 


PHOTOTUBE 
SIGNAL 

e 


p  _L.4E  fsin  <"T    3m3i*T    sinSwT^ 1 

c  ~~  2       If2  [       I  9  25  J 

SIGNAL  WITH  APERTURE  SAME   WIDTH  AS   SCANNING  UN£ 


PHOTOTUBE 
SIGNAL 

e 


9  25 

0.      SIGNAL  WITH  APERTURE  HALF  THE  WIDTH  OF  SCANNING 
LINE 

FIG.  6.    Aperture  Distortion  of  Signals 

when  scanning  such  a  pattern  would  be  50  cycles  per  inch  per  second  of  spot  speed.  If  the 
scanning  line  were  9  in.  long  and  the  drum  speed  100  rpm,  the  fundamental  keying  frequency 
would  be  50  x  9  x  100/60  =  750  cycles  per  second. 

Higher  harmonics  of  this  fundamental  keying  frequency  will  be  present  as  the  subject 
scanned  is  a  square  wave  pattern.  If  the  aperture  of  the  scanner  were  infinitesimal  in 
width  along  the  scanning  line  the  phototube  signal  would  be  a  square  wave  as  ha  Fig.  QB 
and  would  be  very  rich  in  harmonics  as  shown  by  the  Fourier  series  under  this  figure.  This 
perfect  signal  is  never  realized  hi  practice,  nor  is  it  desirable,  for  the  greatly  increased  band 
width  needed  for  transmission  is  not  justified  by  the  small  increase  in  recorded  detail  over 
that  obtained  by  carrying  only  the  fundamental  keying  frequency. 

As  the  aperture  is  made  wider  the  higher  harmonics  become  less  important.  When  the 
aperture  is  the  width  of  the  scanning  line  the  triangular  wave  in  Fig.  6C  is  obtained.  Here 
the  fundamental  is  81  per  cent  and  the  third  harmonic  9  per  cent  as  compared  to  127 
and  42.5  per  cent  for  the  square  wave.  When  the  aperture  is  one-half  the  width  of  the 
scanning  line,  a  condition  normally  used  in  many  scanners,  the  wave  in  Fig.  6D  is  obtained 
where  the  fundamental  is  115  per  cent  and  the  third  harmonic  12.75  per  cent.  When  using 
this  size  of  aperture  very  little  difference  can  be  noticed  in  the  recorded  copy  whether  the 


19-08         FACSIMILE  TRANSMISSION  AND  RECEPTION 

third  harmonic  is  carried  or  suppressed.  For  a  more  complete  analysis  of  scanning  see 
Section  20,  Television. 

Almost  all  facsimile  systems  therefore  carry  only  through  the  fundamental  keying  fre- 
quency as  the  upper  limit  of  the  band  necessary  for  transmission.  In  the  illustration  given 
where  this  fundamental  frequency  was  750  cycles  the  signals  from  the  phototube  would 
have  a  frequency  spectrum  of  0  to  750  cycles  per  second.  This  then  is  the  input  signal  to 
the  scanner  amplifier  system,  and,  as  the  light  reflected  into  the  phototube  is  small,  the 
input  signal  is  usually  very  low  in  amplitude. 

TYPES  OF  SCANNER  AMPLIFIER  SIGNALS.  The  type  of  amplifier  used  in  either 
amplifying  or  processing  the  phototube  signal  will  depend  on  the  types  of  signal  to  be  used 
in  transmission,  and  this  in  turn  will  be  governed  to  some  extent  by  the  transmission  me- 
dium, whether  it  be  wire  line  or  radio.  As  the  lower  frequency  limit  is  zero,  ordinary 
a-c  coupled  audio  amplifiers  cannot  be  used.  In  order  to  carry  this  zero  frequency  or  d-c 
component,  the  phototube  signal  is  usually  caused  to  modulate  a  carrier  wave  either  in 
amplitude  or  frequency.  For  transmission  over  wire  lines  the  carrier  frequency  will  be 
chosen  just  high  enough  to  carry  the  highest  keying  frequency.  For  radio  transmission 
the  radio  carrier  itself  can  be  amplitude  or  frequency  modulated  directly  by  the  facsimile 
signals,  or  a  phone-type  transmitter  may  be  used  and  the  facsimile  modulation  carried  on  an 
audio  subcarrier  as  for  wire  line  transmission.  Finally,  for  short  distances  over  wire  line 
where  no  repeater  stations  or  coupling  transformers  are  used,  a  straight  d-c  amplifier  may 
be  used  between  phototube  and  line.  As  it  is  difficult  to  maintain  drift-free  operation  of  a 
d-c  amplifier  with  sufficient  amplification,  this  last  method  is  seldom  chosen. 

The  signals  transmitted  to  the  recorder  may  thus  be  of  any  one  of  the  following  types: 

1.  Subcarrier  amplitude  modulation  (SCAM). 

2.  Subcarrier  frequency  modulation  (SCFM). 

3.  Radio  carrier  amplitude  modulation, 
a.  Direct  without  subcarrier. 

6.  With  SCAM. 
c.  With  SCFM. 

4.  Radio  carrier  frequency  modulation, 
o.  Direct  without  subcarrier. 

6.  With  SCAM. 
c.  With  SCFM. 

5.  Direct-current  signals. 

For  wire  line  transmission,  signals  of  type  1  or  2  are  commonly  used,  with  5  occasionally 
on  short  control  lines,  for  instance  between  scanner  and  radio  transmitter.  It  is  more  usual, 
however,  if  d-c  signals  are  wanted  to  control  a  radio  transmitter,  to  transmit  signals  of 
types  1  or  2  on  the  control  line  and  detect  to  obtain  the  d-c  signals  for  control  purposes. 

For  radio  transmission  over  long  distances  signals  of  types  3c  or  4a  are  normally  used 
as  they  give  the  most  reliable  results.  For  short  radio  circuits,  as  for  instance  local 
coverage  for  broadcast  facsimile  service,  3c,  46,  or  4c  can  be  used  on  existing  voice  trans- 
mitters. 

Signals  of  types  3a  and  36  are  unreliable  except  for  very  short  distances  and  have  gen- 
erally been  supplanted  by  3c  if  an  a-m  radio  transmitter  is  used. 

TYPES  OF  SCANNER  AMPLIFIERS.  It  can  be  seen  by  the  general  usage  of  the 
various  signals  that  the  two  important  types  of  amplifier  systems  will  be  either  for  am- 
plitude-modulating a  subcarrier  or  frequency-modulating  a  subcarrier.  Where  d-c  signals 
are  required  it  is  customary  to  use  either  one  of  these  and  then  detect  after  amplifying  to 
the  desired  level. 

SUBCARRIER  AMPLITUDE  MODULATION  METHODS.  There  are  three  general 
methods  of  obtaining  signals  of  this  type.  First,  from  the  standpoint  of  the  length  of  time 
it  has  been  in  use,  is  scanning  with  chopped  light.  If  the  light  reaching  the  phototube  is 
made  to  nicker,  either  by  modulating  the  light  itself  or  by  using  a  mechanical  shutter  or 
chopper,  the  output  voltage  developed  by  the  phototube  will  not  be  a  direct  current  but  a 
pulsating  voltage  which  may  readily  be  amplified  by  an  ordinary  a-c  amplifier. 

The  minimum  frequency  of  chopping  will  be  determined  by  the  fundamental  keying 
frequency  of  the  scanning  process;  the  chopper  frequency  will  be  the  carrier  frequency  and 
must  be  high  enough  to  carry  the  shortest  "dot."  In  practice  the  chopper  frequency  is 
usually  made  between  2  and  3  times  the  fundamental  keying  frequency,  with  21/2  a  good 
average.  In  the  previous  example,  with  a  keying  frequency  of  750  cycles  per  second,  a 
carrier  or  chopper  frequency  of  about  1800  cycles  would  be  used.  The  total  bandwidth 
transmitted  would  therefore  be  1800  ±  750,  or  1050  to  2550  cycles.  This  bandwidth  is 
narrow  enough  to  be  carried  on  regular  voice  telephone  circuits  and  is  so  used  for  many  news 
picture  transmissions. 


SCANNER  AMPLIFIERS 


19-09 


CONDENSER 


LAMP 


When  an  ordinary  incandescent  lamp  is  the  light  source,  the  chopper  can  be  either  a  ro- 
tating disk  with  holes  or  slots,  or  a  ribbon  or  reed  vibrating  in  a  magnetic  field.  When  a 
glow-discharge  lamp  is  the  light  source,  such  as  used  in  some  types  of  recorders,  the  light 
can  be  modulated  directly  without  me- 
chanical shutters.  The  same  lamp  can 
thus  serve  for  scanning  or  for  photo- 
graphic recording  on  the  same  machine.  ,  \  /  - 

Two  other  methods  of  obtaining  a-m     /  \  //C-<MICRO SCOPE 

signals  are  shown  in  Figs.  7  and  8.    In     •        -J-       \&^*Z.  r-\64S  PHOTOTUBE 

the  first  the  carrier  is  fed  to  the  special 
phototube  in  a  balanced  bridge  circuit, 
and  a  balance  for  minimum  output 
signal  is  obtained  by  means  of  the  vari- 
able resistor  and  capacitor  with  the 
phototube  dark.  As  light  from  the  sub- 
ject copy  increases,  the  tone  output  of 
the  phototube  increases  and  a  true 
modulation  results.  A  simple  audio 
amplifier  to  build  up  the  modulated 
signals  to  the  required  level  follows  the 
phototube  circuit  shown. 


FIG.  7.    Subcarrier  Amplitude-modulated  (SCAM)  Sig- 
nals Obtained  by  Balanced  Phototube  Circuit 


In  Fig.  8,  the  carrier  is  fed  at  180°  phase  difference  to  the  two  screens  of  a  pair  of  screen- 
grid  tubes.  The  plates  are  connected  together  so  the  outputs  of  the  two  tubes  oppose  each 
other.  By  proper  balancing  of  grid  biases  the  output  tone  may  be  balanced  to  zero  with 
the  phototube  dark.  As  light  to  the  phototube  increases,  the  bridge  is  unbalanced  and  the 
difference  in  output  of  the  two  tubes  is  obtained.  With  this  circuit  it  is  also  possible  to 

balance  for  minimum  signal  with 
maximum  light  for  white  on  the 
phototube,  and  the  output  tone 
will  then  be  a  maximum  for 
black.  Thus  either  positive  or 
negative  modulation  may  be  ob- 
tained by  shift  of  the  balance  ad- 
justments. 

In  either  of  these  circuits*  the 
frequency  of  the  introduced  car- 
rier must  be  high  enough  to  carry 
the  maximum  keying  frequency, 
as  explained  previously  with  the 
light  chopper  systems. 
In  this  type  of  signal  black  is  transmitted  at  one 


FIG.   8. 


Subcarrier  Amplitude-modulated    (SCAM)    Signals 
Obtained  by  Balanced  Modulator 


SUBCARRIER  F-M  METHODS. 

frequency,  white  at  some  different  frequency  either  higher  or  lower  than  that  for  black, 
and  intervening  shades  of  gray  at  proportionate  frequencies  between  these  two  limits.  A 
more  complicated  relationship  exists  between  carrier,  keying  frequency,  frequency  swing, 
and  bandwidth  required  than  with  the  a-m  subcarrier.  This  is,  of  course,  a  true  frequency 
modulation  and  follows  the  same  rules  on  sidebands  as  frequency  modulation  on  a  radio 
carrier.  (See  Section  8  on  frequency  modulation.)  However,  it  has  been  found  that,  with 
a  unity  ratio  of  maximum  keying  frequency  to  the  total  frequency  swing  from  black  to 
white,  the  usable  bandwidth  will  be  confined  to  about  the  same  overall  limits  as  the  a-m 
subcarrier  with  both  upper  and  lower  sidebands.  Again,  as  with  SCAM,  the  lowest  carrier 
frequency  must  be  high  enough  to  carry  the  shortest  dot,  so  the  low  end  of  the  carrier 
swing  should  be  at  least  2  times  the  keying  frequency.  For  the  example  with  a  keying 
frequency  of  750  cycles,  the  carrier  may  swing  from  1500  to  2250  cycles  in  going  from 
white  to  black,  and  the  total  band  spread  (for  all  side  frequencies  greater  than  10  per  cent) 
will  be  from  1125  to  2625  cycles.  The  mid  frequency  for  middle  gray  will  be  1875  cycles, 
and  only  the  first  side  frequency  of  ±  750  cycles  need  be  carried. 

This  is  only  a  deviation  ratio  of  0.5  at  the  highest  keying  frequency,  but  for  all  other  pic- 
ture frequencies  the  ratio  is  higher  and  for  solid  white  backgrounds  is  practically  infinite. 
The  signal-to-noise  improvement  over  SCAM  averages  at  least  12db  for  the  usual  subject 
matter  transmitted.  Still  greater  improvement  in  signal-to-noise  ratio  would  result  from 
increased  swing,  but  at  an  increase  in  bandwidth  that  is  not  justified  in  most  facsimile 
services. 

Two  methods  of  obtaining  this  type  of  signal  are  in  use.  The  first,  shown  in  Fig.  9,  con- 
sists essentially  of  a  beat  oscillator  with  one  of  the  oscillators  being  shifted  over  a  small 
percentage  of  its  frequency  by  a  reactance  tube.  As  the  light  to  the  phototube  increases, 


19-10          FACSIMILE   TRANSMISSION  AND  RECEPTION 

the  frequency  of  the  variable  oscillator  will  be  lowered.  The  two  oscillators  are  set  at 
sufficiently  high  frequencies  so  that  the  reactance  tube  can  readily  swing  the  required 
number  of  cycles  without  changing  amplitude.  For  the  swing  of  1500  cycles  on  white  to 
2250  cycles  on  black  the  fixed  oscillator  could  be  set  at  100,000  cycles  and  the  variable  one 
swung  over  the  range  from  102,250,  with  the  phototube  dark,  to  101,500  cycles  with  the 
phototube  receiving  maximum  light  for  white. 


SCANNER     REACTANCE 
TUBE 


VARIABLE  FIXED 

OSCILLATOR      OSCILLATOR 


DETECTOR 


FILTER 


FiG/9.    Use  of  Reactance  Tube  and  Beat  Oscillator  to  Obtain  Sub  carrier  Frequency-modulated  (SCFM) 

Scanner  Signals 

A  second  method  of  obtaining  SCFM  signals  directly,  without  heterodyning  to  obtain 
the  low  frequencies,  is  shown  in  Fig.  10.  A  resistance-capacitance  oscillator  of  the 
180°  phase  shift  type  is  varied  In  frequency  directly  by  using  a  tube  control  system  as  a 
variable  resistor  in  one  mesh  of  the  phase  shifting  ladder  network.  When  the  control  tube 
has  zero  input  (phototube  dark)  the  bias  is  adjusted  to  set  the  low-frequency  end  of  the 
swing.  As  light  to  the  phototube  increases,  the  tube  resistances  decrease  and  the  frequency 
of  the  oscillator  is  raised.  Input  volume  from  the  phototube  is  adjusted  so  that  the  high- 
frequency  end  of  the  swing  is  just  reached  for  white.  When  the  proper  network  and  tube 
constants  are  chosen,  a  linear  range  as  high  as  2  to  1  in  frequency  may  be  obtained  with 
little  change  in  amplitude. 


OUTPUT 


SCANNER       VARIABLE 
RESISTOR 


RC  OSCILLATOR 
NETWORK 


OSCILLATOR 


FIG.  10.    Use  of  Tube  Control  on  RC  Oscillator  to  Obtain  Sub  carrier  Frequency-modulated  (SCFM) 

Scanner  Signals 

This  circuit,  when  connected  as  shown,  will  give  positive  modulation,  that  is,  an  increase 
in  frequency  for  an  increase  in  light.  To  obtain  negative  modulation  a  reversing  tube 
may  be  connected  between  the  control  grid  and  phototube,  or  the  phototube  may  be 
reversed  and  connected  anode  to  grid,  and  cathode  to  a  negative  supply  potential  below 
ground. 

The  changes  in  frequency  with  this  circuit  are  very  nearly  instantaneous,  because  there 
is  little  stored  energy  in  the  network.  At  the  same  time,  the  frequency  stability  is  adequate 
for  the  purpose. 

It  is  sometimes  necessary  to  use  an  existing  scanning  amplifier  system  having  a-m  output 
and  still  transmit  signals  of  the  SCFM  type.  A  converter  is  then  used  in  which  the  SCAM 
signals  are  rectified  and  filtered  to  obtain  the  original  facsimile  signals,  and  these  are  then 
applied  to  the  control  grid  of  an  SCFM  generator  of  either  of  the  above  types. 


PHOTOGRAPHIC  RECORDING  19-11 

RECORDING  SYSTEMS 

A  perfect  facsimile  recorder  will  build  up  a  copy  of  the  signals  exactly  as  received,  adding 
or  subtracting  nothing,  and  thus  deliver  a  recording  limited  in  detail  only  by  the  scanner 
and  intervening  transmitting  circuit.  The  finished  picture  will  be  almost  identical  in  ap- 
pearance to  the  original  copy. 

Of  the  many  recording  methods,  the  four  most  generally  used  will  be  described  here: 
photographic  recording;  wet  electrolytic  recording;  dry  electrolytic  recording;  and  carbon- 
paper  recording.  Each  of  these  systems  has  advantages  possessed  by  none  of  the  others 
and,  therefore,  will  have  particular  uses  to  which  it  is  the  best  adapted. 

The  length  of  the  scanning  line  and  the  number  of  scanning  lines  per  inch  are  generally 
the  same  as  for  the  scanner,  but  this  agreement  is  not  necessary.  The  recorder  copy  may 
be  made  smaller  or  larger  than  the  original  by  properly  choosing  the  proportions  of  seanning- 
line  length  to  line  advance.  The  product  of  the  total  length  of  the  scanning  line  and  the 
number  of  lines  per  inch  is  called  the  index  of  cooperation;  if  this  value  is  held  constant,  any 
size  recording  may  be  made  with  all  dimensions  correctly  proportioned  to  those  of  the 
original  copy.  Thus,  if  the  scanner  has  a  total  line  length  of  9  in.  and  is  transmitting  at 
100  lines  per  inch,  the  index  of  cooperation  would  be  9  x  100  =  900.  If  it  is  desired  to 
receive  this  picture  on  a  recorder  having  a  scanning-line  length  of  only  4.5  in.,  the  line  ad- 
vance would  be  made  200  lines  per  inch,  and  the  received  copy  would  be  exactly  one-half 
size. 

Other  than  this  index  of  cooperation,  the  other  essential  factor  is  the  number  of  lines 
transmitted  per  minute.  This  is  (often)  termed  "strokes,"  and  the  "per  minute,"  which 
should  be  added,  is  understood.  Thus  the  numbers  40-900  would  signify  a  facsimile  picture 
having  an  index  of  cooperation  of  900  and  transmitted  at  the  rate  of  40  strokes  per  minute, 

4.  PHOTOGRAPHIC  RECORDING 

In  recording  photographically,  the  sensitized  paper  or  film  is  generally  wrapped  on  the 
surface  of  a  drum  and  is  scanned  by  a  small  spot  of  light.  The  light  spot  is  varied  in  in- 
tensity or  size  to  record  the  different  values  of  picture  density.  It  may  be  varied  in  several 
ways — electrically,  mechanically,  or  by  means  of  polarization.  In  the  electrically  varied 
light,  a  neon  or  other  gas  discharge  lamp  is  modulated  in  intensity  by  the  signals.  With  the 
mechanical  system,  the  light  is  steady,  and  varied  either  in  intensity  or  size  of  spot  by- 
means  of  a  vibrating  shutter  or  diaphragm.  In  the  polarized  system,  the  Kerr  cell  is 
interposed  between  the  light  source  and  the  picture  drum,  and  the  light  is  polarized  before 
reaching  the  cell.  The  angle  of  polarization  of  the  cell  is  changed  by  the  picture  signals, 
allowing  more  or  less  light  to  reach  the  picture. 


lm  or  Sensitized  Paper 


FIG.  1.    Photographic  Recorder,  Using  a  Neon  Lamp 

The  first  method  is  more  generally  used  in  this  country,  and  a  simple  recorder  of  this  type 
is  shown  in  Fig.  1.  Here  the  lamp  is  of  the  "point-source"  type.  An  intense  illumination 
is  produced  in  a  small  aperture  within  the  lamp  itself,  and  an  image  of  this  aperture  is 
projected  onto  the  surface  of  the  drum  by  a  lens  system.  The  spacings  of  the  lamp  and  lens 


19-12         FACSIMILE  TRANSMISSION  AND  RECEPTION 

system  are  so  arranged  that  the  aperture  image  is  exactly  the  width  of  a  scanning  line.  If 
several  values  of  line  advance  are  to  be  used  with  the  same  optical  system,  a  variable 
diaphragm  is  introduced  to  regulate  the  size  of  the  image  to  the  proper  value  for  the  line 
width  desired. 

The  relative  motions  of  the  optical  system  and  drum  may  be  any  of  those  used  in  the 
simple  drum  scanner.  Usually  the  drum  is  rotated  while  the  optical  system  is  gradually 
advanced  along  its  surface. 

The  second  method  of  photographic  recording  involves  "valving"  the  amount  of  light 
reaching  the  paper  from  a  steady  light  source,  usually  a  tungsten-filament  lamp.  This  can 
be  done  by  placing  an  oscillograph  mirror  and  aperture  in  the  light  path,  the  position  of  the 
mirror  being  varied  electrically  to  change  the  area  of  the  aperture  exposed.  As  the  amount 
of  light  will  be  directly  proportional  to  area,  a  smooth  variation  of  light  with  signal  is  ob- 
tained. Another  method  consists  of  placing  a  thin  ribbon  in  the  light  path,  just  closing  an 
aperture.  As  the  ribbon  is  twisted  by  the  incoming  signals,  light  is  allowed  to  pass  through 
the  aperture  on  both  sides  of  the  ribbon.  This  system  produces  a  "variable  width  line" 
type  of  recording  similar  in  appearance  to  a  zinc  etching,  or,  if  the  aperture  is  placed  at 
90°  to  the  scanning  direction,  it  will  produce  lines  of  constant  width  but  variable  density. 

In  the  third  method  of  using  polarized  light,  a  Kerr  cell  is  utilized  to  change  the  light 
intensity.  The  optical  system  consists  of  two  Nicol  prisms  placed  between  the  light  source 
and  the  aperture.  These  prisms  are  polarized  in  the  same  plane  and  therefore  pass  light 
through  the  system.  The  Kerr  cell  is  interposed  between  the  prisms,  and  applying  signals 
to  its  polarizing  plates  will  change  its  light-polarizing  properties.  The  amount  of  light 
leaving  the  system  is  therefore  controlled,  and  a  true  modulation  of  the  light  may  be  ob- 
tained. This  system  has  been  used  for  a  number  of  years  in  Europe. 

The  photographic  system  is  far  the  most  accurate  in  its  ability  to  reproduce  completely 
the  signals  received  and  therefore  is  used  in  almost  all  commercial  picture  circuits.  It  has 
one  serious  disadvantage,  however,  in  that  the  received  picture  must  be  developed  before 
the  results  are  known.  The  machine  must  be  loaded  and  operated  in  the  dark.  In  a  fast 
service  this  developing  is  quite  a  handicap,  and  the  fact  that  the  picture  cannot  be  seen  until 
developed  allows  possible  errors  in  the  setting  of  the  equipment  to  go  unnoticed  until  the 
full  time  of  transmission  and  developing  has  elapsed. 

Most  picture  circuits  are  operated  at  speeds  of  6  to  10  square  inches  per  minute,  this 
low  speed  usually  being  due  to  circuit  limitations.  With  adequate  light  and  sensitive  films, 
the  photographic  recorder  is  capable  of  speeds  far  in  excess  of  this  value. 

5.  WET  ELECTROLYTIC  RECORDING 

Electrolytic  recording  is  similar  to  photographic  recording  in  chemical  action  but  has  the 
advantage  of  being  visible  at  once,  or  almost  at  once.  It  may  or  may  not  require  some 
form  of  processing  to  make  the  recording  permanent,  depending  on  the  chemicals  used. 

The  principle  of  operation  is  that  certain  chemicals  turn  very  dark  when  an  electric 
current  is  passed  through  them.  If  a  paper  is  saturated  with  such  a  chemical  and  scanned 
by  a  stylus  contact,  it  may  be  darkened  by  current  at  each  signal  for  black  and  thus  build 
up  the  facsimile  picture. 

The  common  solutions  are  organic  dyes,  though  silver  or  iron  salts  have  sometimes  been 
used  as  in  photography  or  blueprinting.  Some  of  these  solutions  react  very  rapidly  but 
require  high  current  density  to  bring  about  a  dense  enough  black;  others  react  with  much 
less  current  but  require  some  form  of  washing  or  fixing  to  prevent  fading. 

One  recording  solution,  which  gives  a  dense  black  permanent  recording,  used  a  special 
steel  printer  bar  that  is  gradually  worn  away  in  the  recording  process.  This  chemical 
process  is  capable  of  speeds  up  to  50  square  inches  per  minute.  The  chemistry  of  the  color 
formation  is  given  in  U.  S.  patent  2,358,839.  Another  type  of  recording  solution,  which 
uses  a  platinum  bar  that  does  not  take  part  in  the  chemistry  of  recording,  is  given  in  U.  S. 
patent  2,306,471.  The  dye  formed  in  this  case  may  be  any  one  of  several  of  the  azo  dye 
family,  and  is  usually  of  a  deep  purple  color.  Speeds  up  to  160  square  inches  per  minute 
have  been  obtained  with  this  type  of  solution. 

A  machine  using  a  stylus  would  be  a  simple  drum  scanner  with  a  dragging  contact  point 
on  the  surface  of  the  paper.  Another  form  of  electrolytic  recorder  requires  a  continuous 
roll  of  supply  paper  and  prints  one  picture  after  another  without  reloading.  One  form  of 
this  continuous  type  of  recorder  is  shown  in  Fig.  2. 

Here  the  scanning  is  done  by  a  combination  of  a  printer  bar  and  a  helix  on  opposite  sides 
of  the  paper.  The  raised  helix  rotates  at  the  same  speed  as  the  scanner  drum,  thus  making 
one  complete  turn  in  the  length  of  a  scanning  line.  The  point  of  intersection  of  this  helix 
and  bar  will  therefore  travel  across  the  paper  once  for  each  scanning  line.  Current  for 


DRY  ELECTROLYTIC  RECORDING 


19-13 


printing  is  passed  between  the  helix  and  bar,  through  the  paper.  The  bar  is  sprung  slightly 
and  allowed  to  drag  over  the  damp  paper  surface  to  secure  good  contact. 

In  this  machine  the  paper  must  be  moist  to  conduct  the  printer  current  and  allow  the 
chemical  reaction  to  take  place.  For  the  particular  machine  described,  the  paper  is  im- 
pregnated with  the  chemicals  and  kept  at  the  proper  moisture  content  by  storing  in  sealed 
cans.  The  recorder  itself  is  of  moisture-tight  construction  so  that  the  moisture  in  the  paper 
is  retained  until  after  printing. 

In  another  form  of  electrolytic  recorder  a  dry  untreated  paper  is  threaded  through  a 
trough  containing  the  recording  chemicals  before  being  fed  between  the  helix  and  printer 
bar.  After  printing  it  passes  over  a  hot  ironing  roll  to  dry  and  smooth  out  the  recording. 


Paper  feed  roll 
Slot  for  paper  - 
Intersection  point 
Helix 


Motor 
shaft 


Moisture  tight  case 
•  Fixed  printer  bar 

•  Pretreated  damp  paper 
FIG.  2.    Recorder  Using  Wet  Electrolytic  Paper 


The  advantages  of  this  type  of  recorder  are  its  simplicity,  its  visible  recording  feature, 
and  the  extremely  high  speeds  of  which  it  is  capable.  Recording  speeds  as  high  as  80  or  90 
square  inches  per  minute  at  120  lines  per  inch  definition  are  easily  attained,  allowing  a 
full-sized  page  of  8  1/2  by  11  inches  to  be  recorded  in  1  minute.  Though  the  handling  of 
wet  paper  in  the  machine  is  awkward,  this  disadvantage  is  outweighed  in  many  applications 
by  the  speed  of  recording. 


6.  DRY  ELECTROLYTIC  RECORDING 

Several  dry  electrolytic  recording  papers  have  been  developed  for  message  service  fac- 
simile recording:  they  are  much  easier  to  handle  than  the  wet  electrolytic  papers,  though 
not  capable  of  as  high  a  recording  speed.  The  best  known  of  these,  trade-named  Tele- 
deltos,  has  a  light  gray  coating  on  a  dense  black  paper  base  that  has  high  electrical  con- 
ductivity. When  current  is  passed  from  a  small  stylus  through  the  paper  the  metallic 
coating  is  burned  away,  leaving  the  black  paper  underneath  exposed.  The  facsimile  re- 
cording is  built  up  by  scanning  with  a  stylus  and  partially  or  completely  burning  the 
coating  where  gray  or  black  are  to  appear.  See  Fig.  3. 


19-14         FACSIMILE  TRANSMISSION  AND  RECEPTION 

The  advantages  of  this  type  of  recording  are  that  a  permanent  copy  is  produced  with  no 
processing,  the  dry  paper  is  easy  to  handle,  and  the  recorder,  with  only  a  stylus  and  drum, 


is  mechanically  simple.  The  disadvantages  are  that  the  contrast  range  of  the  finished  copy 
is  reduced  by  the  gray  background,  and  the  halftone  scale  is  not  as  linear  as  the  photo- 
graphic or  electrolytic  recorders.  These  do  not  greatly  affect  its  value  for  message  service 
where  the  subject  matter  is  almost  entirely  made  up  of  typing  or  handwriting. 


CARBON-PAPER  RECORDING  19-15 

7.  CARBON-PAPER  RECORDING 

The  first  carbon  recorder  consisted  of  a  stylus  dragging  over  carbon  and  white  papers 
wrapped  on  a^drum.  The  stylus  was  moved  down  to  give  pressure  for  black,  and  lifted  for 
white.  This  is  a  very  simple  form  of  recorder,  but  it  has  the  disadvantage  of  the  photo- 
graphic recorder  in  that  the  picture  is  not  visible  until  the  drum  is  stopped  and  the  carbon 
paper  removed.  It  has  the  advantages  of  cheapness  and  simplicity,  and  the  picture  re- 
quires no  processing  to  be  permanent. 

A  later  form  of  carbon  recorder,  illustrated  in  cross-section  in  Fig.  4,  overcomes  the  dis- 
advantage of  invisible  recording.  Here  the  scanning  is  accomplished  with  a  helix  and 
printer  bar,  as  in  the  continuous  electrolytic  recorder.  Carbon  and  white  paper  are  fed 
between  the  bar  and  helix,  and  after  this  are  separated  so  that  the  surface  of  the  white 


Motor  A  k  /     ,  Intersection  point 


White  papa- 
feed  roU 


Carbon  paper 
take-up  roll 


^  Electro  magnetic  driver 
FIG.  4.    Continuous-feed  Carbon  Recorder 


paper  is  visible  only  a  few  seconds  after  the  printing  process.  The  bar  is  not  allowed  to  drag 
the  paper  but  is  normally  held  away  from  it  by  an  electromagnetic  drive  unit. 

A  signal  for  black  depresses  the  bar,  and  a  black  dot  is  made  by  the  pressure  at  the  inter- 
section of  the  bar  and  helix.  The  carbon  paper  is  drawn  over  guides  and  wound  up  on  a 
take-up  spindle.  The  white  paper  is  fed  by  a  knurled  feed  roll  with  a  series  of  rubber  idlers 
held  against  it,  similar  to  the  paper  feed  of  a  typewriter. 

Only  one  electromagnetic  driver  is  shown  for  the  printer  bar.  However,  if  wide  paper  is 
used,  more  than  one  driver  may  be  necessary  and  the  separate  units  will  be  equally  spaced 
along  the  bar. 

This  method  of  recording  is  very  simple,  uses  cheap  paper,  and  prints  a  very  good  copy  at 
speeds  up  to  10  square  inches  per  minute.  It  is  quite  reliable,  and  the  complete  copy,  with 
no  processing  necessary,  is  visible  only  a  few  seconds  after  recording.  Its  limitations  are 
also  pronounced.  The  printer  bar  is  necessarily  heavier  than  a  stylus,  and  therefore  the 
speed  of  recording  is  limited.  Almost  any  carbon  paper  that  may  be  used  here  will  be  soft 
enough  to  smudge  a  little  when  rubbed  in  the  fingers,  the  same  as  a  carbon  copy  from  a 
typewriter.  More  mechanical  accuracy  is  required  in  building  this  printing  bar  than  in  the 
electrolytic  recorder,  as  the  bar  and  helix  must  be  parallel  to  within  a  few  thousandths  of  an 
inch.  The  depressive  motion  of  the  bar  is  quite  small,  and,  therefore,  a  little  discrepancy  in 
lining  up  the  bar  and  helix  will  result  in  failure  of  part  of  the  paper  to  be  printed  to  a  full 
black.  Damping  of  the  bar  to  eliminate  "bouncing"  and  echo  printing  is  somewhat  of  a 
problem,  but  it  can  be  solved  by  over-powering  the  printing  mechanism  and  absorbing  the 
excess  power  in  a  damping  arrangement  on  the  bar  itself. 

One  advantage  mentioned  separately  here  for  emphasis  is  that  this  type  of  recorder  may 
be  used  to  print  more  than  one  copy  at  a  time.  If  the  printer  bar  action  is  made  sufficiently 
powerful,  several  rolls  of  white  and  carbon  paper  may  be  threaded  into  the  machine  and  a 
number  of  copies  of  the  facsimile  made  at  the  same  time.  As  many  as  8  separate  copies  of  a 


19-16 


FACSIMILE   TRANSMISSION  AND  RECEPTION 


message  have  been  made  experimentally.  Also  the  carbon  paper  may  be  of  the  "hecto- 
graph" type  and  extra  copies  of  the  recording  may  then  be  made  by  the  usual  duplication 
process  of  hectographing. 


8.  COMPARISON  OF  RECORDING  METHODS 

The  recording  method  chosen  for  some  particular  service  will  depend  on  the  quality  of  the 
copy  required,  speed  of  transmission,  cost  of  the  recording  paper,  ease  of  operation,  and 
many  other  factors.  In  a  news  picture  service,  quality  of  the  finished  recording  is  most  im- 
portant, for  the  pictures  are  used  as  masters  to  make  printing  plates.  Photographic  re- 
cording is  therefore  used  for  all  these  pictures,  the  other  factors  being  considered  of  less 
importance.  Operating  speeds  are  maintained  as  high  as  the  wire  lines  or  radio  circuits  will 
permit,  generally  9  or  10  square  inches  per  minute  with  100  lines  per  inch  detail. 

For  message  services,  speed  of  transmission  is  the  most  important,  with  simplified  equip- 
ment able  to  run  unattended  as  a  next  requirement.  With  somewhat  limited  bandwidths 
available  over  wire  lines,  the  high  speed  of  the  wet  electrolytic  process  cannot  be  attained, 
and  the  dry  electrolytic  process  with  its  simpler  recorder*  structure  more  readily  fits  the 
requirements.  Operating  speeds  up  to  30  or  40  square  inches  per  minute  can  be  maintained 
with  this  type  of  recording,  if  the  transmission  band  of  the  line  will  permit.  The  more 
usual  speed  over  most  lines  available  for  this  type  of  service  is  16  to  20  square  inches  per 
minute. 

The  requirements  for  broadcast  facsimile  of  flash  news  services  are  primarily  for  medium 
speed,  direct  printing,  and  a  minimum  paper  cost.  Though  the  wet  electrolytic  recorder 
does  not  use  the  cheapest  paper,  it  meets  the  other  requirements  on  speed  and  direct 
printing.  The  copy  appearance  is  pleasing,  and  the  detail  is  adequate  even  at  higher  speeds 
than  those  now  contemplated.  Such  broadcast  services  will  probably  be  most  effective  at 
speeds  of  around  30  square  inches  per  minute.  Less  than  this  is  too  slow,  and  higher  speeds 
will  tend  to  increase  apparatus  and  paper  costs  too  greatly. 

Carbon  recording  uses  the  cheapest  paper  and  can  run  unattended  for  long  periods  of 
time.  However,  it  is  the  slowest  of  the  recording  methods,  owing  to  the  mechanical  motion 
required  of  the  printer  bar.  Its  highest  operating  speed  at  present  is  about  10  square  inches 
per  minute,  inadequate  for  most  services  but  sufficient  in  some  special  instances  where 
speed  is  less  important. 

9.  RECORDING  AMPLIFIERS 

The  signals  received  for  facsimile  recording  will  usually  be  in  the  form  of  an  a-m  or  f-m 
tone.  This  may  be  either  the  original  SCAM  or  SCFM  tone  transmitted  over  line  or  radio 
by  the  scanner,  or  it  may  be  one  obtained  by  heterodyning  in  a  radio  transmission  of  direct 
frequency  modulation.  If  of  the  SCAM  type,  the  signals  will  be  applied  directly  to  one  of 
the  recording  amplifiers  described  later.  If  of  the  SCFM  type  the  signals  must  first  be 
changed  into  an  equivalent  SCAM  signal  by  limiting  and  then  passing  the  constant- 
amplitude  signal  through  a  slope  demodulating  filter. 

Such  a  converter  unit  is  shown  in  Fig.  5.  The  incoming  SCFM  signal  presumably  will 
have  spurious  amplitude  modulation  superimposed  by  fading  if  received  by  radio  or 


LIMITER         FILTER  DRIVER 


LOW  OR  HIGH   PASS 
SLOPE    FILTERS 


FILTER  CURVES 


FIG.  5.    Limiter  Amplifier  and  Slope  Demodulating  Filters  for  SCFM-type  Signals 

changes  in  line  transmission  characteristics  if  received  by  wire  line-  Therefore  it  is  first 
passed  through  a  limiter  amplifier  of  several  stages,  so  that  the  limiter  output  signal  will,  be 
of  constant  amplitude  over  wide  changes  in  input  level.  This  constant-amplitude  signal 
then  goes  through  either  a  low-  or  a  high-pass  filter,  or  both  if  pushpull  output  is  desired. 


RECORDING  AMPLIFIERS 


19-17 


These  filters  are  designed  to  have  slow,  straight  cut  off  slopes  so  that  the  amplitude  of  the 
output  signals  will  vary  linearly  with  frequency. 

In  the  circuit  shown  the  two  end  frequencies  of  the  SCFM  swing  are  labeled  /i  and  /2  for 
the  white  and  black  frequencies.  If  demodulated  in  the  upper  filter  the  output  will  be 
greatest  for  /i  and  least  for  /a,  while  the  output  of  the  lower  filter  is  the  opposite.  For  un- 
distorted  demodulation  these  filter  slopes  must  be  linear  beyond  the  band  from  /i  to  /2  by 


SCAM  ^  §    o 


RECORDER 


+  B 


THRESHOLD 

FIG.  6.    Printer  Amplifier  for  Maximum  Output  Current  with  Minimum  Signal  Amplitude 

the  amount  of  the  side  frequency  spread  of  the  SCFM  wave.  For  the  previous  example  of  a 
swing  of  1500  cycles  for  white  to  2250  cycles  for  black,  and  band  spread  from  1125  to 
2625  cycles,  the  filter  slope  must  be  linear  over  the  entire  range  from  1125  to  2625  cycles. 

The  output  of  either  slope  filter  will  be  the  equivalent  of  an  SCAM  signal  in  amplitude 
envelope  and  can  be  used  in  the  printer  amplifier  in  the  same  manner. 

Four  simplified  diagrams  of  printer  amplifiers  to  actuate  the  various  types  of  recorders 
are  shown  in  Figs.  6,  7,  8,  and  9.  The  first  two  are  suitable  for  photographic  recorders  using 
a  glow  discharge  lamp  to  expose  the  film  or  paper.  In  each  the  SCAM  signal  is  rectified  and 


RECORDER 

"  RECTIFIED  ^ 

'  THRESHOLD 

FIG.  7.    Printer  Amplifier  for  Maximum  Output  Current  with  Maximum  Signal  Amplitude 


filtered  by  a  capacitor  across  the  volume  control  to  obtain  the  facsimile  signals.  The  output 
tube  is  then  either  driven  to  lower  output  current  as  signal  amplitude  increases,  as  in  Fig.  6, 
or  to  higher  output  current  as  amplitude  increases,  as  in  Fig.  7.  The  particular  one  used 
will  depend  on  the  direction  of  modulation  of  the  signal,  and  whether  the  drum  is  loaded 
with  film  to  make  a  negative  or  with  photographic  paper  to  make  a  positive  copy. 

For  recorders  using  a  pushpull  magnetiqally  driven  printer,  or  driver  units  of  a  carbon 
recorder,  these  two  types  of  amplifiers  may  be  combined,  as  in  Fig.  8.  As  signal  amplitude 
increases,  the  upper  output  tube  will  be  driven  to  lower  current  and  the  lower  tube  to  higher 

MIRROR 
GALVANOMETER 


SCAM^g 
INPUT 


RECTIFIERS    VOLU 


RECORDER 


FIG.  8.     Push-pull  Printer  Amplifier  for  Magnetically  Driven  Printer  Systems 

current.  When  properly  adjusted  true  pushpull  output  is  obtained,  and  the  sum  of  the 
two  output  tube  currents  will  be  constant.  This  constant-sum  current  can  be  used  as  shown 
in  the  cathode  return  bias  resistor  to  furnish  the  threshold  bias  required  by  the  lower 
output  tube. 

For  electrolytic  recorders  of  either  the  wet  or  dry  type  it  is  a  good  safety  measure  to  run 
the  helix  or  recorder  drum  at  ground  potential.  To  accomplish  this  an  amplifier  such  as  in 
Fig.  9  may  be  used.  A  pushpull  power  amplifier  at  cut-off  bias  amplifies  the  SCAM  signals 


19-18         FACSIMILE  TRANSMISSION  AND  RECEPTION 

to  a  power  level  sufficient  for  printing.  This  a-c  signal  is  applied  directly  between  the 
ground  drum  and  stylus  for  dry  electrolytic  recording,  but  it  must  be  rectified  and  applied 
between  helix  and  bar  in  the  proper  polarity  for  most  wet  electrolytic  printing  processes.  In 


VWV— i 


RECORDER 


FIG.  9.     Printer  Amplifier  for  Either'Polarity  of  Electrolytic-type  Recording.    With  non-polarity  sensi- 
tive papers,  such  as  Teledeltos,  the  a-c  output  is  used  without  rectification. 

some  of  these  the  color  forms  on  the  anode  side,  in  which  case  the  bar  is  made  positive  with 
respect  to  the  helix;  in  others  the  color  forms  on  the  cathode  side  of  the  paper,  and  so 
the  bar  is  made  negative. 


SYNCHRONIZING  AND  PHASING 


10.  SYNCHRONIZING 

In  every  facsimile  system,  it  is  necessary  that  the  recorder  follow  the  scanner  over  the 
paper  in  order  to  produce  an  undistorted  recording.    The  principle  of  synchronizing  may  be 

better  understood  by  refer- 


ring to  Fig.  1.  For  clarity 
the  picture  elements  are 
shown  much  larger  in  pro- 
portion than  they  really  are. 
As  the  scanner  starts  the 
picture  on  element  1,  the 
recorder  also  starts  on  its 
element  1.  As  succeeding 
scanning  lines  are  drawn,  the 


Scanner 
FIG.  1. 


Recorder 
Synchronism  of  a  Facsimile  System 


recorder  must  follow  exactly,  or  the  copy  will  be  distorted  by  a  misplacing  of  the  elements. 
Besides  having  the  synchronizing  correct,  the  recorder  must  be  in  "phase"  with  the 
scanner,  as  illustrated  in  Fig.  2.  Even  though  the  two  drums  are  rotating  at  exactly  the 
same  speed,  if  they  are  not  in  phase  the  border  of  the  picture  will  be  misplaced.  The  re- 
corder drum  must  start  each  scanning  line  at  the  same  time  the  scanner  is  starting  that 
scanning  line,  or,  as  shown  in  Fig.  2C,  the  border  will  be  somewhere  between  the  two  edges  of 
the  paper  instead  of  being  exactly  divided.  Phasing  and  synchronizing  become  the  same 
problem  only  if  the  phasing  line  of  the  picture,  or  border,  controls  the  speed  of  the  recorder. 
Where  the  synchronizing  frequency  is  independent  of  the  "phasing  line,"  or  of  a  much 
higher  frequency  than  that  of  the  phasing  line,  the  two  problems  are  separate  and  must  be 
treated  separately.  This  is  generally  the  case  in  most  commercial  systems  in  use  today. 


A.  Original  in  scanner  B.  Recorder  in  phase  C.  Recorder  out  of  phase 

FIG.  2.    Phasing  of  a  Facsimile  Recorder 

Before  going  into  the  means  of  synchronizing  and  phasing,  the  effect  of  imperfect  syn- 
chronizing should  be  shown,  to  illustrate  the  problem  better.  Figure  3  shows  the  effect  of 
an  error  in  synchronizing  on  a  unidirectional  scanning  system  and  on  a  back-and-forth 
scanner.  The  error  illustrated  here  is  that  the  recorder  is  running  faster  than  the  scanner 
by  a  very  small  percentage.  In  scanning  the  vertical  line  in  A,  the  recorder  gets  farther 
along  its  scanning  line  each  time,  moving  the  recorded  line  farther  and  farther  to  the  right. 


SYNCHRONIZING 


19-19 


The  result,  in  a  unidirectional  system,  is  shown  in  B.  In  a  back-and-forth  scanning  system, 
the  result  is  much  more  pronounced,  alternate  lines  moving  apart,  as  in  C.  The  result,  for 
the  recorder  being  too  slow,  would  appear  the  same  with  this  method  of  scanning,  while 
with  the  unidirectional  system  the  line  would  have  slanted  down  to  the  left  instead  of  down 
to  the  right. 

The  accuracy  of  the  synchronizing  may  vary  with  the  particular  system.    In  commercial 
work,  the  necessary  accuracy  is  very  high.    In  a  system  scanning  at  60  strokes,  100  lines  per 


A.  Original  in  scanner 


B.  Recorder  fast,  unidirec- 
tional scanning 


[C.  Recorder  fast,  "back- 
and-forth"  scanning 


FIG.  3.    Recorder  Not  Perfectly  Synchronized 

inch,  and  each  line  of  9-in.  length,  the  total  length  of  scanning  line  per  vertical  inch  of 
paper  is  900  in.  In  a  picture  of  10-in.  length,  this  total  scanning  line  length  will  then  be 
9000  in.  A  good  copy  will  be  made  if  the  total  drift  in  the  border  of  the  picture  is  not  over 
1/4  in.  in  this  10-in.  length  of  picture.  Thus,  the  synchronizing  system  must  hold  an  ac- 
curacy of  1/4  part  in  9000,  or  1  part  in  36,000.  It  must  hold  this  rate  for  the  whole  trans- 
mitting time  of  nearly  17  minutes.  Actually  most  commercial  systems  have  synchronizing 
equipment  accurate  to  1  part  in  100,000  or  better. 

TUNING-FORK  FREQUENCY  STANDARD.  In  short-distance  facsimile  transmis- 
sion, as  for  instance  local  coverage  of  a  broadcast  facsimile  service,  the  same  a-c  power 
supply  is  often  available  for  both  scanner  and  recorders.  Synchronism  is  then  simplified  by 
driving  both  scanner  and  recorders  with  ordinary  synchronous  motors  connected  to  the 
common  supply.  In  long-distance  transmission,  or  across  the  sea,  this  is  not  possible,  and 
synchronism  is  generally  maintained  by  controlling  the  motors  of  both  scanner  and  recorder 
by  accurate  frequency  standards.  Such  frequency  standards  usually  take  the  form  of  very 
accurate  tuning  forks  that  will  hold  a  constant  frequency  to  within  1  part  in  100,000  or 
better.  Crystal  standards  could  also  be  used,  but,  as  the  control  frequency  for  the  motor 
is  usually  low,  less  dividing  of  frequency  is  required  with  a  fork. 

To  hold  the  required  accuracy,  the  fork  is  usually  held  at  a  fixed  temperature  by  thermo- 
stat control,  or  a  temperature-compensated  bimetallic  type  of  fork  is  used.  Figure  4  shows 
one  method  of  driving  a  tuning  fork  by  using  it  as  a  resonant  coupling  circuit  in  a  vacuum- 
tube  oscillator.  By  changing  the  driving  power  supplied  to  the  fork,  a  vernier  control  on 


I  FREQUENCY 
S-?YERN1ER 


POWER   AMPLIFIER 

OR 
THYRATRON    INVERTER 


FIG.  4.    Fork  Control  of  Synchronous  Motor 

its  frequency  is  obtained.  The  fork  frequency  may  be  amplified  by  tubes  or  thyratron  in- 
verters to  a  power  level  sufficient  to  drive  a  synchronous  motor  directly,  or  it  may  be  used 
in  other  ways  to  control  motor  speed. 

In  one  application,  an  1800-cycle  tuning  fork  is  used  both  to  supply  carrier  tone  for  an 
SCAM  signal  and  is  also  amplified  to  a  power  of  about  10  watts  to  drive  an  1800-cycle 
synchronous  motor.  The  synchronous  motor  is  brought  up  to  speed  by  a  d-c  motor,  as  it  is 
not  self  starting.  In  applications  where  a  standard  60-cycle  synchronous  motor  is  used,  the 
fork  frequency  can  be  made  60  cycles,  or  divided  down  to  60  cycles  from  some  higher 
frequency. 

MAGNETIC  BRAKE  SYNCHRONIZING.  Another  type  of  control  circuit  is  shown 
in  Fig.  5.  An  induction  motor,  or  other  type  of  motor  having  good  speed  regulation,  is 
used  to  drive  the  recorder  or  scanner  and  is  controlled  to  an  exact  speed  by  means  of  a 


19-20          FACSIMILE   TRANSMISSION  AND   RECEPTION 


RECORDER      TONE 
(OR  SCANNER)  GENERATOR! 

II5V  60** 

FIG.  5.    Magnetic  Brake  Synchronizing  System 


magnetic  brake.  A  tone  generator  of  the  phonic  wheel  type  is  mounted  on  the  motor  shaft, 
having  the  correct  number  of  poles  to  generate  a  frequency  equal  to  the  fork  frequency 
at  the  correct  motor  speed.  A  phase  comparison  between  this  generated  frequency  and  the 

fork  control  frequency  is 
then  used  to  vary  the 
brake  current  and  make 
the  system  lock  into  syn- 
chronism with  the  fork. 

A  wave  analysis  of  the 
brake  action  is  shown  in 
Fig.  6.  The  generated  tone 
and  the  fork  tone  are  each 
amplified  and  limited  to 
give  the  two  square  waves 
e\  and  e%,  as  in  Fig.  6a. 
These  two  waves  are  added 
together  algebraically  in 
the  mixer  tube  to  give  the 
waves  shown  in  Fig.  66  and 
are  full-wave  rectified  to 
give  the  waves  in  Fig.  6c. 
The  result  is  then  the  grid 
voltage  of  the  tubes  sup- 
plying the  braking  current, 
shown  in  Fig.  Qd. 

Two  conditions  are  illus- 
trated. The  waves  on  the 
left  side  show  the  motor  leading  the  fork  by  a  small  phase  angle,  and  the  pulses  of  brake 
current  are  of  full  amplitude  but  narrow  in  time,  so  that  the  average  brake  current  is 
small.  If  the  motor  tries  to  speed  up  for  any  reason,  such  as  an  increase  in  line  voltage  or 
lightening  of  the  mechanical  load,  it 
will  advance  in  phase  with  respect 
to  the  fork.  The  waves  on  the  right 
side  illustrate  how  the  brake  current 
pulses  are  increased  in  time  width  to 
give  a  higher  average  brake  current 
that  tends  to  slow  down  the  motor. 
The  brake  is  thus  turned  full  on,  or 
full  off,  by  the  square  pulses,  and  the 
correct  average  is  obtained  mechani- 
cally rather  than  by  smoothing  these 
pulses  in  a  filter.  Hunting  is  thereby 
almost  completely  eliminated.  As 
the  two  waves  are  squared  up  before 
comparing,  the  change  in  ratio  of 
time  on  to  time  off  is  a  linear  func- 
tion with  phase-angle  changes  of  zero 
to  180°. 

As  very  effective  brake  action  can 
be  obtained  by  passing  the  pulsating 
direct  current  through  the  windings 
of  an  ordinary  induction  motor,  a 
special  design  of  brake  is  not  neces- 
sarily required.  This  system  is  es- 
pecially well  adapted  for  very  high- 
speed facsimile  where  motor  powers 
as  high  as  1/e  hp  are  needed  and  con- 


n 
l 

-i— 

/"FC 

RK  e, 

V  MOTOR      ] 

ft 

A'lr'* 

! 
T~ 

j 

i~" 
i 

\ 

l 

-4 

1 
i 

-+ 

LIMITER   OUTPUTS,  e  AND  Ce 


MIXER  OUTPUT,  (e,*e z) 


rjDDUT    TTTF 


C.      RECTIFIED     SUM    OF 


FIG.  6.    Brake  Operation  as  Motor  Goes  from  Small  Phase 
Lead  on  Fork  to  Large  Lead 


d.  BRAKE  CURRENT 

trol  must  be  at  high  frequency  to 
limit  phase  displacement  with 
changes  in  load  or  line  voltage. 

START-STOP  SYNCHRONIZATION.  The  first  methods  used  for  facsimile  synchro- 
nization were  generally  of  the  start-stop  type,  and,  although  such  a  synchronizing  system 
is  now  practically  obsolete  for  facsimile,  it  is  still  used  on  some  forms  of  automatic  tape 
printers,  such  as  the  teletype. 

In  start-stop  systems,  the  scanner  is  generally  operated  at  a  constant  speed  and  has  a 


PHASING 


19-21 


"phasing  line"  of  a  considerable  time  length.  During  this  phasing  line  interval,  the  recorder 
will  have  finished  its  scanning  line  and  stopped  automatically.  The  scanner  sends  a  pulse 
at  the  start  of  the  succeeding  scanning  line,  and  a  clutch,  or  similar  mechanical  apparatus, 
starts  the  recorder  on  the  next  scanning  line.  A  governor-controlled  motor,  or  some  other 
fairly  accurate  drive,  is  used  to  maintain  the  recorder  at  a  constant  speed  for  the  duration 
of  each  scanning  line. 

The  chief  merit  of  this  system  is  that  the  errors  in  speed  of  the  recorder  are  not  accumu- 
lated, each  scanning  line  starting  afresh.  The  greatest  possible  discrepancy  in  synchron- 
izing, therefore,  is  the  error  in  any  one  scanning  line  itself,  and  this  can  be  made  quite  small. 
The  disadvantage  is  that  the  mechanics  of  such  a  system  must  be  quite  complicated,  and  a 
definite  starting  pulse  must  be  received  or  the  entire  scanning  line  is  lost.  The  speed  of  the 
entire  system  must,  therefore,  be  quite  slow  to  insure  that  these  two  factors  do  not  interfere 
with  the  picture.  A  complicated  scanning  system  cannot  be  started  instantaneously  at  a 
high  scanning  speed,  as  allowances  must  be  made  for  the  inertia.  Fading  of  the  signal,  if 
received  by  radio,  would  cause  such  a  system  completely  to  miss  whole  scanning  lines  if 
starting  pulses  were  not  received. 

For  use  by  line,  such  a  system  has  advantages,  as  an  ordinary  governor  will  synchronize  a 
motor  accurately  enough  for  the  purpose,  and  failure  to  receive  a  starting  pulse  is  rare. 

OTHER  SYNCHRONIZING  SYSTEMS.  Certain  recording  systems  require  no  syn- 
chronizing at  all,  and  such  methods,  sometimes  used  for  cable  transmission  of  pictures, 
involve  setting  up  a  certain  number  of  picture  elements  by  machine,  or  by  hand,  and  sending 
a  tape  of  this  series  of  elements  in  numerical  order.  The  recording  is  then  assembled  by 
hand,  usually  requiring  a  competent  artist  to  give  the  picture  a  lifelike  appearance.  This 
method  has  been  used  for  a  number  of  years  with  great  success  over  wire  and  cable.  The 
Bartholemew-McFarlane  system  or,  in  shorter  terms,  the  "Bartlane"  system  is  a  variation 
of  this  method. 

The  synchronizing  frequency  of  the  scanner  is  sometimes  sent  over  the  radio  or  wire  line, 
and  an  amplifier  is  used  to  build  this  signal  up  to  a  value  where  it  is  able  to  drive  or  control  a 
synchronous  motor  on  the  recorder.  Such  methods  are  satisfactory  on  line  transmissions 
and  short  radio  circuits  but  cannot  be  depended  on  for  long  radio  transmissions. 


11.  PHASING 

The  phasing  of  the  recorder  to  the  incoming  signals  can  be  accomplished  either  manually 
or  automatically.  The  simplest  manual  method  is  to  throw  the  recorder  out  of  syn- 
chronism and  let  it  drift  until  some  indicator,  such  as  a  neon  lamp  fed  by  the  phase  signal 
pulses,  indicates  in  phase,  and  then  to  re-establish  synchronism  to  hold  this  position. 

In  a  simple  automatic  system,  used  on  many  news  photo  equipments,  the  recording  drum 
is  driven  through  a  clutch,  and  a  projecting  ear  on  the  drum  is  arranged  to  engage  against  a 
stop-pin  having  a  magnetic  release.  At  the  start  of  the  picture  the  drum  is  held  in  start 
position  by  this  pin,  and  the  clutch  slips.  When  the  start  phase  signal  is  received  the  drum 
is  released  and,  being  of  low  mechanical  inertia,  starts  rotating  almost  immediately  and  in 
phase.  The  stop-pin  is  locked  out  automatically  when  tripped,  so  the  drum  continues  to 
rotate  for  the  duration  of  the  picture  transmission  time.  A  phasing  system  suitable  for  a 
continuous  recorder  is  shown  in  Fig.  7.  A  phase  signal  is  transmitted  at  the  start  of  each 

COMMUTATOR 
CIRCUIT   CLOSE£ 
EXCEPT     DURING^ 
PHASE    POSITION 


MOTOR  SUPPLY 
POWER 


PHASE    SIGNAL  •*• 
DC    PULSE    INPUT 


FIG.  7.    Automatic  Phasing  of  Continuous  Recorder 

scanning  line,  and  this  signal  is  selected  out  of  the  picture  signals  by  frequency  or  amplitude 
discrimination. 

It  is  rectified  and  passed  to  the  input  circuit  as  a  d-c  pulse  at  the  start  of  each  line.    In 
series  with  this  input  is  a  commutator  that  is  closed  at  all  times  except  for  a  short  gap  at  the 


19-22         FACSIMILE  TRANSMISSION  AND  RECEPTION 

correct  phase  position.  When  the  recorder  is  running  in  phase,  this  gap  opens  the  circuit 
for  a  slightly  longer  time  than  the  duration  of  the  phase  pulse,  so  the  tube  receives  no 
signal.  If  the  pulse  arrives  at  any  other  time,  it  passes  through  the  closed  portion  of  the 
commutator  and  causes  the  tube  to  draw  a  pulse  of  current  to  operate  the  relay.  This 
opens  the  motor  circuit  momentarily  and  causes  it  to  drop  synchronism.  To  insure  the 
relays  staying  open  long  enough  for  the  motor  to  lose  V2  or  1  cycle  of  synchronism,  the  con- 
tacts B  of  the  relay  connect  the  capacitor  from  relay  coil  to  ground,  and  the  charge  current  of 
the  capacitor  holds  the  relay  in  operating  position  for  a  fixed  length  of  time.  The  resistor 
across  this  capacitor  is  too  high  to  pass  enough  current  to  hold  the  relay  in,  but  it  bleeds  the 
capacitor  to  zero  charge  between  pulses.  The  motor  is  thus  jogged  out  of  synchronism 
once  for  each  pulse  received  in  an  out-of-phase  position,  and  this  process  continues  until  the 
correct  phase  position  is  reached,  and  the  commutator  again  opens  the  pulse  circuit  at  the 
correct  time. 

TRANSMISSION  CHARACTERISTICS 

12.  WIRE  LINE  TRANSMISSION 

Transmission  of  facsimile  signals  by  either  wire  line  or  radio  puts  more  exacting  require- 
ments on  the  circuit  than  telephone  or  telegraph  transmission.  This  is  largely  due  to  the 
exact  timing  of  the  signals,  which  requires  that  delay  equalization  of  a  wire  line  be  much 
more  precise  than  for  telephone  work.  Any  appreciable  difference  in  arrival  time  at  the 
recorder  of  the  high-  and  low-frequency  components  of  the  picture  will  show  as  transients 
and  ghosts  that  exaggerate  the  outlines  of  objects  in  the  picture  and  may  even  make  typing 
unreadable.  Accurate  delay  equalization  is  therefore  required  on  all  but  short  lines. 

The  amount  of  delay  equalizing  necessary  is  also  directly  affected  by  the  speed  of  the 
transmission  and  by  the  bandwidth  required.  For  the  previous  example  of  a  maximum 
keying  speed  of  750  cycles,  the  shortest  dot  to  be  transmitted  is  1/1500  second,  or  0.667 
millisecond.  Any  difference  in  delay  equalization  over  the  band  of  1800  db  750  cycles,  or 
from  1050  to  2550  cycles,  should  not  exceed  a  fraction  of  this  0.667  millisecond  or  noticeable 
distortion  will  result.  In  this  case,  the  line  should  be  delay-equalized  to  =t  0.25  millisecond 
over  the  1050-  to  2550-cycle  bandwidth.  In  many  long  lines  used  for  facsimile,  this  maxi- 
mum delay  error  of  db  0.25  millisecond,  over  a  band  of  1000  to  2600  cycles,  is  maintained. 

If  the  speed  of  transmission  were  doubled,  the  maximum  permissible  delay  error  would  be 
halved  and  the  bandwidth  doubled  at  the  same  time.  The  problem  of  getting  good  enough 
lines  is  therefore  increasingly  difficult  as  speed  is  increased. 

This  exactness  of  delay  equalization  can  be  compared  to  regular  voice  circuits  where  10  or 
even  more  milliseconds'  delay  difference  does  not  appreciably  affect  quality  of  speech. 

Where  a-m  signals  are  being  used,  the  line  must  also  be  equalized  for  amplitude  over  the 
transmission  band,  but  this  is  usually  an  easier  problem.  Where  SCFM  signals  are  used  the 
amplitude  characteristic  of  the  line  is  relatively  unimportant,  but  the  delay  characteristic 
must  be  as  good  as  for  SCAM  signals. 

For  short  transmissions  of  less  than  100  miles  portable  scanners  are  sometimes  operated 
into  ordinary  coin-box  phones  and  over  regular  long-distance  lines.  This  practice  is  satis- 
factory in  some  cases,  but  distortion  is  much  greater  than  over  the  specially  equalized  lines, 
and  picture  quality  is  therefore  lower. 

13.  RADIO  TRANSMISSION 

Radio  transmission  is  beset  with  more  difficulties  than  line  transmission  and,  for  long 
distances,  is  generally  slower.  Many  factors  enter  in  long-distance  radio  transmissions, 
such  as  fading,  multipath  delays,  interference,  echos,  and  other  forms  of  distortion  that 
must  be  corrected  for,  or  else  the  speed  must  be  decreased  until  the  particular  distortion 
present  is  reduced  sufficiently  to  be  no  longer  objectionable. 

Rapid  changes  in  transmission  distance,  due  to  varying  heights  of  the  ionized  layers, 
give  the  effect  of  varying  delay  times  on  wire  lines  and  are  the  limiting  factor  on  speed  of 
transmission.  Some  of  the  most  pronounced  effects  can  be  eliminated  by  suitable  directive 
antennas  with  limited  pick-up  angles,  to  eliminate  the  next  higher  order  of  skip  or  hop. 
When  such  antennas  are  used,  and  the  proper  choice  of  carrier  frequency  made  for  the  dis- 
tance, speeds  up  to  10  sq  in.  per  minute  are  generally  possible  on  circuits  as  long  as  New 
York  to  London,  and  a  speed  of  6  sq  in.  per  minute  is  very  reliable. 

In  earlier  radio  facsimile  systems,  dot-halftoning  was  developed  so  that  a  keyed  on-off 
CW  type  of  transmission  might  be  used.  Limiting  the  incoming  signal  then  allowed  most  of 


TAPE  FACSIMILE  19-23 

the  results  of  fading  to  be  removed.  However,  all  present  long-distance  transmissions  are 
made  either  by  using  SCFM  on  regular  voice  transmitters,  or  direct  frequency  modulation, 
or  frequency  shift  of  the  radio  carrier.  The  dot-halftone  systems  have  therefore  become 
obsolete,  and  at  the  same  time  speeds  have  increased  from  2  to  3  times  that  possible  with  the 
dot  systems.  With  either  of  these  newer  methods  limiting  can  be  done  either  at  radio  fre- 
quency for  frequency  modulation  or  at  audio  frequency  for  SCFM  and  fading  can  be 
largely  removed. 

For  short  distances,  or  when  using  ultra-high-frequency  relaying,  the  speed  is  not  so  lim- 
ited, for  multipath  troubles  do  not  enter,  and  wide  bands  may  be  used. 


SPECIALIZED  APPLICATIONS 

14.  DUPLICATORS 

Many  applications  of  facsimile  have  been  made  that  illustrate  that  it  is  not  limited  solely 
to  the  transmission  of  pictures  over  long  distances.  The  scanner  and  recorder  can  be 
mounted  on  the  same  shaft  and  a  duplicator,  or  copying  machine,  obtained.  Two  types  of 
facsimile  duplicators  are  in  use,  each  having  definite  advantages  over  other  forms  of  du- 
plicators in  certain  applications. 

In  one  type  of  duplicator  a  wet  electrolytic  recorder  is  combined  with  a  rapid-loading 
type  of  scanner,  such  as  shown  in  either  Fig.  4,  p.  5,  or  Fig.  5,  p.  6.  The  speed  of  opera- 
tion is  very  high,  85  sq  in.  per  minute,  and  a  full-sized  letter  page  8  1/2  by  11  in.  is  copied  in 
slightly  over  a  minute,  with  120  lines  per  inch  detail.  The  wet  paper  passes  over  an  ironing 
roll  after  printing,  and  a  finished  dry  copy  is  thus  delivered. 

As  both  scanner  and  recorder  are  rigidly  coupled  to  the  same  driving  motor,  no  synchro- 
nizing or  phasing  is  required.  The  amplifier  system  from  phototube  to  printer  becomes  very 
simple,  as  there  is  no  transmission  and  reception  problem  over  line  or  radio. 

This  type  of  duplicator  is  useful  where  only  a  few  copies  each  of  a  large  number  of 
originals  are  required.  As  reflected  light  is  used  the  original  may  be  opaque,  entirely  un- 
suitable for  blueprinting.  It  thus  compares  with  photo-copying  (though  the  cost  of  the 
printing  paper  is  less)  but  eliminates  the  necessity  of  developing  and  fixing. 

MULTIFAX.  In  Multifax,  a  master  mimeograph  stencil  is  cut  by  facsimile  methods 
so  that  a  large  number  of  copies  of  the  original  may  be  made.  Copy  containing  illustrations 
and  diagrams  that  would  be  almost  impossible  to  make  up  by  ordinary  means  can  thus  be 
obtained  in  quantity,  without  going  through  more  expensive  printing  processes. 

The  scanner  and  recorder  are  on  the  same  shaft,  as  in  Duplifax,  to  eliminate  the  need  of 
synchronizing  and  phasing.  The  stencil  is  clamped  on  a  recording  drum  and  cut  by  a  stylus 
which  is  vibrated  at  high  frequency.  While  vibrating,  the  stylus  is  moved  towards  the 
drum  for  black  and  delivers  a  large  number  of  blows  to  the  stencil  to  displace  the  wax.  The 
stylus  is  retracted  for  white  so  that  it  just  misses  touching  the  stencil.  The  vibrating  stylus 
has  no  tendency  to  drag  out  or  tear  the  stencil,  as  it  would  if  pressure  only  were  applied,  and 
so  much  finer  detail  can  be  realized  than  with  hand-cut  stencils. 

A  full  letter-sized  master  stencil  can  be  prepared  from  the  original  in  10  or  15  minutes  by 
this  method,  about  the  time  required  for  typing  a  stencil  without  illustrations.  Another  ad- 
vantage is  that  the  original  is  prepared  on  white  paper,  and  printed  diagrams  or  illustra- 
tions may  be  pasted  in  place  without  being  hand  drawn  on  the  stencil. 

15.  TAPE  FACSIMILE 

Tape  facsimile  systems  are  a  special  adaptation  of  facsimile  in  which  the  recording  is  printed 
on  a  narrow  slip  similar  to  that  used  in  news  tickers.  The  scanning  may  be  optical  or  me- 
chanical, but  the  recorders  for  either  type  of  scanning  are  usually  of  the  helix  and  printer- 
bar  type,  somewhat  like  a  miniature  version  of  the  carbon  recorder. 

Where  optical  scanning  is  used,  the  message  is  typed,  or  handwritten,  on  the  trans- 
mitting tape,  and  this  original  copy  then  fed  through  a  scanner  similar  to  that  shown  in 
Fig.  1.  A  spot  of  light  is  traversed  across  the  width  of  the  tape  by  the  combination  of  a 
rotating  prism  and  a  fixed  system  of  three  cylindrical  lenses  and  one  right-angle  prism. 
One  stroke  across  the  tape  is  obtained  for  each  face  of  the  prism  that  passes.  These  scan- 
ning lines  are  very  short,  usually  1/4  to  1/2  in.  The  phototube  signals  generated  by  the  re- 
flected light  are  then  used  to  obtain  either  SCAM  or  SCFM  signals  in  the  same  manner  as 
in  any  page  facsimile  scanner  amplifier. 


19-24         FACSIMILE  TRANSMISSION  AND  RECEPTION 


For  mechanical  scanning  the  transmitter  takes  a  form  similar  to  a  tape-teletype  machine, 
in  which  each  letter  is  represented  by  a  disk  similar  to  an  "Omnigraph"  disk.  Each  disk 
is  a  commutator  and  keys  a  series  of  signals  that  will  form  a  facsimile  image  of  that 'char- 
acter when  printed  by  the  recorder.  Operation  of  this  form  of  tape  scanner  is  then  similar 


'Phototube 


Scanning  prism 
Pickup  mirrors   /rotating  600  rpm 


Cylindrical  lenses 


Prism 
Cylindrical  lens 


FIG.  1.    Tape  Facsimile  Scanner 

to  that  of  a  teletype  machine,  but  with  the  output  signals  coded  for  facsimile  recording 
rather  than  for  operating  a  typewriter  type  of  receiver. 

Either  type  of  scanner  will  operate  the  tape  recorder  shown  diagrammatically  in  Fig.  2. 
The  helix  is  very  small  for  a  scanning  line  as  narrow  as  this,  and  it  can  be  inked  directly  by 
the  ink  roller  shown  instead  of  by  using  a  slip  of  carbon  paper  to  supply  the  coloring  matter 
as  in  the  page  type  of  recorders.  Otherwise  the  recorder  action  is  exactly  the  same  as  for 
the  page  carbon  recorder  shown  in  Fig.  4,  p.  15.  Tape  systems  are  built  for  message  service 

Felt  inking 
roller 


Helix  drum 
600  rpm 


Electromagnetic 
driver 


FIG.  2.    Tape  Facsimile  Recorder 


and  do  not  need  to  have  the  fine  detail  usually  required  in  a  picture  system.  Whether  op- 
tical or  mechanical  scanning  is  used,  the  characters  transmitted  are  large  block  type  to  re- 
duce keying  frequency  and  to  get  as  large  a  number  of  words  per  minute  as  possible  in  a 
narrow  channel. 


BIBLIOGRAPHY  19-25 

The  scanning-line  length  is  very  little  greater  than  the  height  of  a  letter,  so  that  there  is 
little  waste  time  in  margins.  For  instance,  if  3/i6-in.  block  type  is  used,  the  scanning  line 
will  be  about  1/4  in.  long.  To  transmit  this  type,  the  shortest  dot  necessary  would  be  about 
0.020  in.,  and  a  maximum  number  of  cycles  per  scanning  line  would  be  6.  With  the  con- 
stants of  60  lines  per  inch  and  60  lines  per  second,  1  in.  of  tape  per  second  is  transmitted 
with  a  keying  speed  of  not  over  360  cycles  per  second.  This  tape  speed  of  1  in.  per  sec  will 
represent  about  60  words  per  minute. 

While  this  band  of  360  cycles  for  60  words  per  minute  is  about  8  1/2  times  that  required 
for  the  commercial  automatic  7  unit  codes,  where  42  cycles  (or  bauds)  per  second  represents 
60  words  per  minute,  this  disadvantage  is  largely  offset  in  instances  where  signals  are  poor, 
or  noise  high.  With  tape  facsimile,  interference  can  obliterate  a  letter,  but  it  cannot  make 
it  print  a  wrong  one. 

BIBLIOGRAPHY 

Standards  on  Facsimile,  Definition  of  Terms,  1942,  Supplement  of  Proc.  I.R.E.,  Vol.  30,  No.  7,  Part  IV. 

Radio  Facsimile,  RCA  Institutes  Technical  Press,  1938,  pp.  112  to  128  contains  a  very  complete  bib- 
liography of  facsimile  up  to  October  1938. 

Mathes  and  Whitaker,  Radio  Facsimile  by  Subcarrier  Frequency  Modulation,  RCA  Rev.,  October 
1939,  pp.  131-154. 

Artzt,  M,,  Frequency  Modulation  of  RC  Oscillators,  Proc.  I.R.E.,  July  1944,  pp.  409-414. 

Bliss,  W.  H.,  Subcarrier  Frequency  Modulation,  Proc.  I.R.K,  August  1943,  pp.  419-423. 

Felch,  E.  P.,  Measuring  Delay  on  Picture  Transmission  Circuits,  Bell  Lab.  Rec.,  January  1936,  pp. 
154-157. 

Hinshaw,  F.  A.,  Delay  Equalizers  for  Telephotograph  Transmission,  Bell  Lab.  Rec.,  February  1936, 
pp.  193-197. 

Mertz,  P.,  The  Telephotograph  Line,  Bell  Lab.  Rec.,  February  1936,  pp.  178-184. 

Mertz  and  Pfleger,  Irregularities  in  Broad-Band  Wire  Transmission  Circuits,  Bell  Syst,  Tech.  J.,  Vol. 
16,  pp.  541-559  (October  1937). 

Schulman,  D.,  Facsimile  Synchronizing  Methods,  Electronics,  March  1946,  pp.  131-133. 

Wise  and  Coggeshall,  The  Handling  of  Telegrams  in  Facsimile,  Proc.  I.R.E.,  May  1941,  pp.  237-242. 

U,  S.  Patent  2,358,839,  Iron  Bar  Wet  Electrolytic  Recording. 

U.  S.  Patent  2,306,471,  Axo  Dye  Wet  Electrolytic  Recording. 


SECTION  20 
TELEVISION 


PRINCIPLES  AND  THEORY 


1.  Physiological  Requirements  ...........  02 

2.  Subdivision  of  Picture;  Effect  of  Scan- 

ning Rates  ........................  03 

3.  Resolution   and  Flicker   Requirements; 

Band   Width  ......................  06 

4.  Pick-up  Devices  ....  .................  07 

5.  Picture  Display  Devices  ..............  08 

6.  Synchronizing  .......................  11 

7.  The  Video  Signal  ....................  13 

8.  The  Composite  Signal  ................  16 

9.  The  Radio-frequency  Signal  ...........  17 

10.  Standards  ...............  .  ...........  20 

TELEVISION  BROADCASTING 

BY  T.  J.  BtrzALSKi,  A.  L.  HAMMERSCHMIDT,  AND 

F.  J.  SOMEBS 

11.  Lens  Aperture  Required  ..............  21 

12.  Studio  Camera  Design  ................  22 

13.  Studio  Equipment  ...................  26 

14.  Gamma  (Transfer  Characteristic)  ......  29 

15.  Aperture  Correction  ..................  30 

16.  Film  Pick-up  ........................  31 

17.  Master  Control  Position  ..............  32 

18.  Pulse  Measurements  .................  33 

19.  Overall  Video  System  Response  ........  35 

20.  Television  Field  Pick-up  Equipment  ----  36 


ABT.  PAGE 

21.  Relay  of  Television  Signal 37 

22.  Transmitter  Plant  Terminal  Equipment  41 

TELEVISION  RECEIVERS 
BY  W.  F.  BAILEY  AND  R.  J.  BBUNN 

23.  Antennas 47 

24.  R-f  Circuits 47 

25.  Modulator  and  Local  Oscillator 49 

26.  Picture  I-f  Amplifier 49 

27.  Picture  Channel  Second  Detector 52 

28.  Video  Amplifiers  and  Display 53 

29.  Noise  Limiters 55 

30.  Sound  Amplifiers 56 

31.  Synchronization 57 

32.  Scanning 59 

33.  Power  Supply 62 

OTHER  FORMS  OF  TELEVISION 

BY  A.  V.  LOTJGHREN 

34.  Television  Standards  of  Foreign  Coun- 

tries   64 

35.  Theater  Television 64 

36.  Color  Television 65 

37.  Binocular  Television 67 

38.  Television  for  Special  Services 67 

39.  Diplexing  of  Picture  and  Sound 67 


20-01 


TELEVISION 


PRINCIPLES  AND  THEORY 

By  A.  V.  Loughren 

Television  is  defined  as  "the  electrical  transmission  and  reception  of  transient  visual 
images."  * 

Television  technique  for  monocular,  monochrome  pictures  has  developed  sufficiently  to 
lead  to  adoption  by  the  Federal  Communications  Commission  of  standards  for  broad- 
casting. 

1.  PHYSIOLOGICAL  REQUIREMENTS 

The  performance  required  of  a  television  system  is  determined  by  physiological  require- 
ments which  must  be  met  for  the  performance  to  be  acceptable.  Section  14  discusses 
these  in  detail.  They  vary  from  individual  to  individual,  but  generally  acceptable  design 
values  for  the  several  quantities  have  been  arrived  at  based  on  extensive  tests.  These 
requirements  include : 

Resolution.  An  observer  with  good  eyesight  is  able  to  resolve  successive  contrasting 
objects  individually  subtending  as  little  as  1  minute  of  arc.  A  square  subtending  1  minute 
of  arc  on  a  side  corresponds  to  a  solid  angle  of  approximately  10~7  steradian.  (Refer- 
ences 1,  2,  3.) 

Field  of  View..  A  normal  eye  is  capable  instantaneously  of  critically  observing  a  field 
of  the  order  of  0.001  steradian.  Since  the  eye  direction  can  be  quickly  and  readily  changed, 
a  much  greater  field  than  this  is  available  within  a  very  short  interval  of  time.  For  sus- 
tained viewing  of  images  the  viewing  distance  of  four  to  eight  times  the  picture  height 
chosen  by  most  observers  produces  an  image  field  of  the  order  of  0.02  to  0.07  steradian. 
Such  a  field  is  200,000  to  700,000  times  the  minimum  resolvable  solid  angle  (reference  3). 

Sharpness.  Sharpness  is  the  subjective  quantity  corresponding  to  the  objective  quan- 
tity * 'resolution."  Figure  1  shows  the  relation  between  sharpness  and  resolution;  it  in- 


Relative  Subject  Sharpness  in  Laminal  Units 

5  i  A  0  *  »  R 

_-  — 

—  •  ' 

— 

,^-«« 

^ 

^^ 

/ 

/ 

/ 

/ 

/ 

-16 

/T 

ousands 

of  Rgure 

;  of  Confi 

ision  in  t 

ie  Conve 

ntional  F 

ield  of  VK 

w 

3                20               40              60               80             100             12 

1  •  I          1          1                    ! 

0               140              160               10 

43           2          1.5                     1.0                          0 

7                           as 

Area  of  Figures  of  Conf us ion-Ste radians  xlCT6 
FIG.  1.     Sharpness  vs  Resolution  (from  Baldwin,  Ref.  4) 

dicates  that  increasing  the  resolution  by  making  the  size  of  the  figure  of  confusion  less  than 
1.5  X  10  ~6  steradian  increases  the  sharpness  only  slightly  (reference  4). 


*RMA. 


20-02 


SUBDIVISION  OF  PICTTJKE 


20-03 


Brightness.  Because  of  its  essentially  logarithmic  response  and  its  ability  to  control 
admitted  light  by  means  of  the  iris  opening,  the  human  eye  is  capable  of  observing  objects 
whose  brightnesses  lie  within  the  range  from  4  X  10 ~5  to  4000  ft-lamberts.  It  is  found, 
however,  that  satisfactory  viewing  requires  restriction  of  this  range.  Under  conditions 
of  low  ambient  illumination,  highlight  brightnesses  as  low  as  1  ft-lambert  are  found  ac- 
ceptable; however,  under  conditions  of  normal  artificial  and  natural  lighting  indoors,  high- 
light brightnesses  as  great  as  200  ft-lamberts  are  desirable.  Values  of  10  to  100  ft-lam- 
berts are  suitable  for  design  purposes. 

Contrast.  The  total  contrast  range  instantaneously  perceptible  to  the  eye  is  believed 
to  be  about  40,000  :  1.  However,  reproductions  exhibit  contrast  ranges  from  10  :  1  for 
rather  unsatisfactory  images  to  200  :  1  for  the  best  photographic  transparencies.  Tele- 
vision pictures  having  a  contrast  range  of  30  :  1  have  been  judged  reasonably  satisfactory. 

Color.     See  Section  14  and  references  5  and  25. 

Depth.     See  Section  14. 

Moving  Objects.  Ideally,  reproduction  of  a  picture  of  a  moving  object  requires  that 
each  elementary  area  of  the  picture  change  synchronously  with  the  corresponding  changes 
in  the  original  scene  caused  by  the  motion  of  the  object.  It  is  known,  however,  that  the 
resolution  of  the  eye  for  moving  objects  is  much  poorer  than  for  stationary  objects.  It  is 
consequently  permissible  to  reproduce  the  picture  at  finite  intervals  rather  than  con- 
tinuously. For  most  purposes  the  interval  of  1/24  sec  is  short  enough  to  leave  with  the 
observer  the  illusion  that  motion  is  continuous  rather  than  discontinuous. 

Shape  and  Size  of  Picture.  A  rectangular  shape  with  the  width  equal  to  four-thirds 
of  the  height  has  been  found  generally  acceptable.  This  ratio  is  defined  as  the  aspect  ratio. 

The  minimum  acceptable  size  for  reproduced  pictures  is  believed  to  lie  in  the  range 
between  4  by  5.33  in.  and  7  1/2  by  10  in.  Smaller  pictures  produce  fatigue  within  a  short 
time  unless  special  devices  are  worn  by  the  viewer.  The  maximum  acceptable  picture 
size  is  determined  primarily  by  the  viewing  distance  available.  For  household  use  pic- 
tures up  to  15  by  20  in.  are  suitable,  while  for  use  in  halls  and  theaters  much  larger  ones 
are  appropriate. 


2.  SUBDIVISION  OF  PICTURE;  EFFECT  OF  SCANNING  RATES 

Methods  of  transmitting  and  reproducing  a  television  picture  fall  into  two  classes. 
Both  classes  depend  on  the  subdivision  of  the  scene  into  a  sufficiently  large  number  of 
elementary  areas.  In  one  class  of  system  a  separate  transmission  channel  is  provided 
continuously  for  each  element.  The  large  number  of  elements  required  for  a  satisfactory 
picture  has  shown  systems  of  this  class  to  be  impracticable.  In  the  other  class,  which  all 
present  practical  systems  use,  the  elements  are  connected  to  a  single  transmission  chan- 
nel successively  in  an  ordered  sequence  common  to  both  transmitter  and  receiver.  This 
process  is  called  scanning. 

SCANNING.  An  area  to  be  scanned  is  subdivided  into  elements  each  of  which  is 
connected  to  the  transmission  channel  periodically  in  some  regular  sequence.  In  its 
simplest  form  the  operation  consists  in  starting  at 
the  upper  left-hand  corner  of  the  picture,  traversing 


successively  the  row  of  elements  along  the  top  of  the 
picture  from  left  to  right,  following  with  a  similar 
traverse  one  element  width  lower,  and  continuing 
this  process  until  the  bottom  of  the  scanned  area  is 
reached.  A  single  traverse  across  the  picture  in  one 
direction  is  called  a  scanning  line.  The  complete 
scanning  pattern  is  referred  to  as  a  raster.  This 
process  is  shown  diagrammatically,  for  a  picture 
containing  only  a  few  scanning  lines,  in  Fig.  2. 
The  electrical  frequency  components  produced  by 
thus  scanning  a  fixed  image  may  be  shown  to  con- 
sist of:  (1)  a  d-c  component;  (2)  components  at  the 
vertical  scanning  frequency  and  its  harmonics;  (3) 


FIG.  2.     Simple  Raster 


components  at  the  horizontal  scanning  frequency  and  its  harmonics;  (4)  components  at 
sum  and  difference  frequencies  of  the  above.  Physically  the  d-c  component  represents 
the  average  brightness  of  the  image.  The  components  at  the  vertical  frequency  and 
its  harmonics  represent  bands  extending  horizontally  across  the  picture.  The  com- 
ponents at  the  line  frequency  and  its  harmonics  represent  vertical  bands.  The  compo- 
nents at  sum  and  difference  frequencies  represent  inclined  bands.  If  the  picture  changes 
with  time,  side  bands  are  added  to  some  or  all  of  these  components  (references  6  and  29) . 


20-04 


TELEVISION 


Vertical  Resolution.  The  scanning  spot  in  the  usual  case  is  not  rectangular  nor  does  it 
exhibit  uniform  effectiveness  over  its  area.  As  a  typical  example  of  this,  Fig.  3  shows  the 
distribution  of  light  intensity  over  the  scanning  spot  of  a  cathode-ray  tube.  If  the  spot 
moves  rapidly  in  one  direction,  forming  a  line,  its  effective  distribution  in  the  other  direc- 
tion assumes  some  such  form  as  that  shown  in  Fig.  4  at  A.  This  figure  also  illustrates, 
at  B,  the  condition  under  which  a  flat  field  of  illumination  is  produced  by  the  overlap  of 
successive  lines,  and  at  C  a  line-to-line  spacing  which  fails  to  produce  a  flat  field.  Similar 
relations  exist  with  respect  to  the  scanning  spot  in  the  photosensitive  device  at  the 
transmitter. 

The  photo  device  at  the  transmitter  should  be  adjusted  to  satisfy  the  flat-field  criterion. 
Failure  to  do  this  results  in  a  type  of  distortion  known  as  "beads,"  which  is  not  susceptible 
of  any  subsequent  correction. 

Vertical  width  of  confusion,  defined  as  the  average  width  in  the  reproduced  image  of  a 
very  narrow  line  appearing  before  the  transmitter,  positioned  at  a  slight  angle  with  re- 
spect to  the  scanning  lines,  is  equal  to 
V2  times  the  scanning-line  pitch  (ref- 
erence 7). 

Horizontal  Resolution.  If  the  scan- 
ning spot  of  a  pick-up  device  moves 
horizontally  across  a  vertical  line  of 
negligible  width,  the  resulting  electri- 
cal impulse  describes  the  horizontal 
characteristic  of  the  spot.  Typical 
forms  of  this  impulse  are  shown  in 
Fig.  5.  The  width  of  a  rectangle  hav- 
ing the  same  maximum  height,  and 
including  the  same  area  as  the  spot 
characteristic  (as  shown  dotted  in 
Fig.  56),  is  an  approximate  measure 
of  the  duration  of  the  impulse.  Since 
the  spot  is  traveling  at  a  fixed  velocity, 
the  abscissa  in  Fig.  5  represents  not 
only  time  but  also  distance.  It  may, 
for  analysis,  be  transformed  into 
steady-state  amplitude  and  phase 
characteristics  as  functions  of  fre- 
quency (for  example,  by  means  of  the 


J' 

1 


I 

I 


8.0 


Distance  Alpng  Diameter—  mm 

FIG.  3,     Distribution  of  Intensity  across  Scanning  Spot 
(After  Zworykin,  Proc,  IJ2.2?.,  Dec.,  1933) 


Fourier  integral  theorem) .  The  effec- 
tive band  width  may  be  expressed 
approximately  by  the  width  of  a  rectangular  area  having  the  same  maximum  height 
and  same  included  area  as  the  frequency  characteristic.  It  may  be  shown  that  to  a  useful 
approximation  the  effective  spot  width  in  seconds  (or  microseconds)  is  related  to  the  ef- 
fective band  width  in  cycles  (or  megacycles)  by  the  equation : 


_ 
v. 


(i) 


where  t  is  the  time-duration  of  the  equivalent  rectangular  electrical  transient  and  fc  is  the 
cutoff  frequency  of  the  equivalent  rectangular  frequency  characteristic.  A  signal  gen- 
erated by  a  scanning  spot  moving  across  a  narrow  line  as  just  described,  transmitted  elec- 
trically and  reproduced  by  a  reproducing  device,  will  have  its  frequency  spectrum  modified 
both  by  the  electrical  circuits  and  by  the  equivalent  transmission  characteristic  of  the 
spot  of  the  reproducing  device.  These  several  characteristics  may  be  multiplied  together 
to  produce  the  overall  transmission  characteristic  of  the  system.  The  corresponding  ef- 
fective horizontal  width  of  confusion  is  then  obtained  by  applying  to  this  characteristic 
the  inverse  transformation  by  means  of  the  Fourier  integral  theorem.  As  an  approxima- 
tion, the  effects  of  several  sources  of  limitation  on  the  frequency  band  width,  connected  in 
cascade,  on  the  effective  overall  band  width,  and  on  the  duration  of  the  shortest  repro- 
ducible impulse  are  given  by  the  equations 

(2) 


and 


fc  VT^V 


t  •*  vV  +  fe2  4-  - 


(3) 


SUBDIVISION  OF  PICTURE 


20-05 


o 


232 


So  long  as  the  frequency  characteristic^ 
or  the  transient  impulse  form  reason- 
ably well  approximates  the  error  func- 
tion 


the  relations  given  above  are  close  ap- 
proximations (references  7  and  8)  . 

The  scanning  spot  is  usually  symmet- 
rical and  thus  exhibits  no  phase  distor- 
tion. The  electrical  circuits,  however, 
are  potential  sources  of  phase  distortion. 
Phase  distortion  affecting  low-frequency 
components  of  the  reproduced  picture 
tends  to  alter  the  vertical  or  lateral 
shading  of  the  picture.  Phase  distortion 
affecting  high-frequency  components- 
tends  to  give  the  picture  a  "bas  relief" 
effect  in  which  edges  of  objects  in  the- 
image  may  be  preceded  or  followed  by 
bright  or  dark  outlines.  Sharp  cutoff 
in  the  amplitude  characteristic  produces 
"overshoots"  superficially  similar  to 
high-frequency  phase  distortion.  The 
two  effects  differ  in  that  if  a  symmetrical 
object  is  scanned  the  distortion  due  to 
sharp  amplitude  cutoff  will  be  itself' 
symmetrical,  whereas  that  due  to  phase- 
distortion  will  be  opposite  in  its  char- 
acter on  the  two  sides  of  the  image  (ref- 
erence 9). 

FLICKER.  Flicker  is  not  inherent  in 
television.  It  is  a  consequence  of  the 
use  of  scanning  in  conjunction  with 
picture  display  devices  in  which  energy 
is  supplied  to  a  given  elementary  area 
of  the  display  device  for  only  a  minute 
fraction  of  the  total  time.  The  use  of 
such  display  devices  with  the  scaiining. 
sequence  of  Fig.  2.  at  a  picture  repeti- 


a.  Rectangular 
Uniform  Spot 


r 


b.  Circular 
Shaded  Spot 


by  interlaced  scanning. 


FIG.  5.    Electrical  Impulse  Produced  by  Scan- 
ning a  Narrow  Vertical  Line 

tion  rate  of  25  or  30  per  sec,  produces 
severe  flicker  at  brightnesses  even  below 
1  ft-lambert.  Restriction  of  the  maxi- 
mum brightness  to  this  level  is  not  ac- 
ceptable- 

^  A  major  improvement  is   obtained 

In  this  form  of  scanning,  shown  in  Fig.  6,  the  vertical  component 


£ 

CO 


20-06 


TELEVISION 


of  velocity  of  the  scanning  spot  is  doubled  as  compared  to  that  of  Fig.  2  so  that  while  the 
spot  crosses  the  picture  from  left  to  right  it  falls  an  interval  equal  to  twice  the  distance 
between  scanning  lines.  If,  therefore,  the  spot  starts  at  the  top  center  to  trace  line  1 
after  completing  line  1  it  goes  to  the  left-hand  side  of  the  picture  and  starts  line  3,  con- 
tinuing in  this  manner  until  it  reaches  the  bottom  right  side  of  the  picture.  At  this'point 
the  vertical  retrace  takes  place  and  the  spot  returns  to  the  top  left  of  the  picture  and 
starts  to  scan  line  2,  followed  by  lines  4,  6,  etc.,  until  the  lower  center  of  the  picture  is 
reached.  (For  purposes  of  illustration  the  vertical  retrace  time  is  assumed  to  be  very 

small.)  This  method  of  operation  has  the  conse- 
quence that  for  an  observer  at  such  a  distance  that 
he  just  fails  to  resolve  individual  scanning  lines  the 
effective  flicker  frequency  has  been  doubled;  thus, 
very  much  greater  brightness  is  permissible  without 
any  increase  in  the  picture  repetition  rate.  This 
advantage  is  only  slightly  impaired  when  the  ob- 
server's distance  is  such  that  he  can  commence  to 
resolve  individual  scanning  lines.  The  analogy  be- 
tween the  practice  of  interlacing  and  the  motion- 
picture  practice  of  interrupting  the  light  at  a  rate 
greater  than  the  frame  frequency  should  be  noted. 

In  interlaced  scanning  the  time  required  by  a  single 
vertical  traverse  of  the  picture  is  no  longer  equal  to 
the  time  required  to  scan  a  complete  picture.  A  single  vertical  scan  is  called  a  field; 
a  complete  picture  is  called  a  frame.  The  customary  variety  of  interlace  thus  has  two 
fields  per  frame.  Higher  orders  of  interlace  have  been  proposed  but  have  not  found 
widespread  use. 

It  is  desirable  that  the  scanning  processes,  both  horizontal  and  vertical,  repeat  exactly 
from  cycle  to  cycle.  If  the  number  of  lines  to  a  complete  frame  is  an  odd  number,  and  a 
frame  consists  of  two  fields,  correct  interlace  and  uniformity  of  repetition  of  the  scanning 
operation  go  hand  in  hand.  This  arrangement  is  called  "odd-line  interlace." 

3.  RESOLUTION  AND  FLICKER  REQUIREMENTS;  BAND  WIDTH 

In  article  2  it  was  noted  that  the  vertical  width  of  confusion  was  equal  to  V2  times  the 
scanning-line  pitch;  hence 


FIG.  6.     Interlaced  Scanning 


where  V  is  the  picture  height  and  n  the  number  of  useful  scanning  lines  per  picture. 
was  also  noted  that  the  horizontal  width  of  confusion,  in  seconds,  was 


It 

(1) 

(5) 

Wh  =  2fc  (6) 

It  is  desirable  that  the  horizontal  and  vertical  widths  of  confusion  be  approximately  equal. 
Thus 


Since 

Wh  =  tv 
where  v  is  the  horizontal  spot  velocity,  it  follows  from  eq.  (1)  that 


whence 


fc 


2\/27 


(8) 


The  number  of  useful  lines  in  the  complete  picture,  n,  will  be  less  than  the  total  number 
of  line-periods  in  the  picture  time-interval,  nf,  by  a  factor  a  (usually  0.90  to  0.93)  introduced 
to  provide  time  for  the  vertical  return  of  the  spot.  The  line-repetition  rate  or  line- 
scanning  frequency,  in  cycles  per  second,  is  the  product  of  the  picture-repetition  fre- 
quency N  and  the  total  number  of  lines  per  picture,  n\ 

If  no  time  were  allowed  for  horizontal  return  of  the  scanning  spot  from  the  right  to  the 
left  of  the  raster,  the  horizontal  velocity  would  have  the  value 

v  =  WNn'  (9) 


PICK-UP   DEVICES  20-07 

where  W  is  the  picture  width.  Retrace  time  must  be  provided,  thus  reducing  the  useful 
portion  of  the  line-period  from  unity  to  a  fraction  b,  usually  0.82  to  0.85.  The  actual 
velocity  is  then 

WNn' 

*  -  -g-  (io) 

Substituting  this  value  in  eq.  (8) 

_  nn'NW       an^N       W 

fe   —  7^  —  ^~    X    ~^  (11) 

2\/2bV       2V26       V 
and,  since  W/V,  the  aspect  ratio,  is  4/3, 

fc  =  ~^-  n*N  (12) 


If  the  picture  is  viewed  from  a  distance  equal  to  four  times  its  height,  the  angular  width 
of  confusion  is  found  from  eq.  (4)  to  be 

rrr  V2 

W*  =  J^7    radian  (13) 

while  the  corresponding  square  solid  angle  is 

1 
S  ~  — ^~~^>     steradian  (14) 

In  article  1  it  was  stated  that  the  figure  of  confusion  should  not  exceed  1.5  X  10"6 
steradian.  However,  the  method  of  defining  the  boundaries  of  the  figure,  in  the  study 
there  referred  to  (reference  4) ,  differed  from  that  of  article  2  sufficiently  to  require  intro- 
duction of  the  factor  1/1.9  when  this  figure  is  applied  to  the  preceding  analysis.  Using 
the  resulting  value  of  0.8  X  10  ~*  steradian,  and  choosing  the  value  0.90  for  a,  eq.  (14) 
yields  the  value 

ri  —  439  lines  per  picture 

The  American  television  standards  have  been  set  at  525  lines  per  picture,  thus  more  than 
meeting  the  resolution  requirement  in  the  vertical  direction. 

The  vertical  repetition  rate  must  be  at  least  25  and  preferably  30  per  second  for  the  com- 
plete picture  in  order  to  minimize  flicker,  even  with  interlaced  scanning.  It  is  advantageous 
to  make  the  rate  an  integral  submultiple  of  the  power  supply  frequency,  to  eliminate  dis- 
turbance of  interlace  by  stray  fields;  a  vertical  rate  of  30  per  second  is  therefore  chosen. 
The  corresponding  line  frequency  is  30  times  525,  or  15,750  cycles  per  second. 

The  effective  cutoff  frequency,  as  defined  in  article  2,  is  obtained  by  substituting  in 
eq.  (12)  the  values  a  =  0.90,  b  =  0.84,  nf  =  525,  N  =  30.  Thus 

fe  =  0.505n'2JV  =  4.1S  Me 

With  a  frequency  band  including  components  up  to  4  or  4.5  Me,  the  gradual  cutoff  re- 
quired to  avoid  severe  "overshoots"  (and  usually  obtained  automatically  as  a  consequence 
of  scanning  spot  distributions)  results  in  an  effective  band  width  which  rarely  exceeds  3 
Me  (reference  7).  The  condition  of  equal  vertical  and  horizontal  widths  of  confusion 
(eq.  [71)  is  thus  not  usually  obtained;  moderate  departures  from  this  condition  are  known 
to  be  of  only  minor  importance  (reference  4) . 

4.  PICK-UP  DEVICES 

Early  work  on  television  employed  mechanical  scanning  of  the  object  or  scene  to  be 
televised.  The  optical  system  focused  an  image  on  a  plane  at  which  a  disk  provided  with 
apertures  spaced  about  its  periphery  was  interposed.  A  single  rotation  of  the  disk  pro- 
duced one  complete  scan  of  the  image.  The  photoresponsive  device  located  behind  the 
disk  responded  instantaneously  to  the  light  transmitted  by  the  apertures  as  they  suc- 
cessively traversed  the  image.  In  a  modification  of  this  arrangement  the  scanning  process 
was  applied  to  the  light  which  illuminated  the  object.  This  modification  reduced  the 
total  amount  of  light  incident  on  the  subject,  and  the  accompanying  heat,  very  con- 
siderably. 

Analysis  of  these  mechanical  scanning  methods  shows  that  the  light  incident  upon  the 
photoresponsive  device  is  inversely  proportional  to  the  number  of  picture  elements,  and 
that,  for  an  acceptable  number  of  elements  in  the  picture,  it  is  not  practicable  to  increase 
the  scene  lighting  and  the  pick-up  lens  size  sufficiently  to  produce  a  useful  signal-to-noise 
ratio  (reference  26).  The  limitation  inherent  in  mechanical  pick-up  systems  encouraged 


20-08  TELEVISION 

the  development  of  photo  responsive  devices  capable  of  storing  light  energy  over  the 
entire  scanning  period,  thus  permitting  an  improvement  of  several  orders  of  magnitude 
in  the  signal-to-noise  ratio.  These  devices  are  described  in  Section  15. 


5.  PICTURE  DISPLAY  DEVICES 

Early  picture  display  devices  employed  a  light  source  of  instantaneously  controllable 
intensity  (such  as  a  crater-type  glow-discharge  lamp  or  a  high-intensity  arc  whose  light 
output  was  modulated  by  a  Kerr  cell)  in  conjunction  with  a  mechanical  scanning  device 
similar  to  that  described  in  connection  with  the  preceding  article.  With  arrangements 
of  this  sort  each  picture  element  is  illuminated  for  a  time  interval  corresponding  to  its 
own  duration. 

The  average  illumination  of  a  picture  element  at  the  viewing  screen  is  equal  to  the  il- 
lumination intensity  during  the  picture  element  divided  by  the  ratio  of  frame  duration  to 
picture  element  duration.  A  typical  value  for  this  ratio  is  300,000.  In  consequence  of 
this  factor,  and  the  limited  intrinsic  brightnesses  of  convenient  light  sources,  mechanical 
scanning  systems  as  heretofore  proposed  have  been  largely  supplanted  by  electronic  scan- 
ning using  cathode-ray  tubes  (reference  2) . 

In  a  modification  of  a  mechanically  scanned  television  reproducer,  Scophony  developed 
a  method  of  storing  the  light-modulation  information  in  a  liquid  cell  in  such  fashion  that 
an  individual  picture  element  could  be  illuminated  for  a  period  many  times  the  element's 
own  duration.  Practical  difficulties  have  prevented  widespread  use  of  this  arrangement 
(references  10,  11,  12,  and  13). 

In  a  cathode-ray  tube  the  instantaneous  power  concentration  in  the  scanning  spot  may 
reach  values  of  10  to  1000  kw  per  sq  in.  Efficiencies  of  fluorescent  materials  in  converting 
from  electrical  to  luminous  energy  lie  in  the  range  of  5  to  10  per  cent.  On  this  basis, 
average  highlight  brightnesses  of  a  few  thousand  foot-lamberts  are  possible.  The  cathode- 
ray  tube  has,  therefore,  become  the  accepted  television  picture-reproducing  device.  Cath- 
ode-ray tubes  are  described  more  fully  in  Section  15. 

Electronic  control  of  the  opacity  or  alternatively  the  light-reflection  coefficient  of  a 
surface  has  been  employed  for  certain  special  purposes.  In  one  form  the  effect  of  an  in- 
jected electron  in  producing  opacity  in  a  transparent  alkali  halide  crystal  is  employed. 
In  another  arrangement  a  layer  of  flakelike  particles  such  as  graphite,  suspended  in  a 
fluid,  is  used;  in  the  absence  of  electric  field  the  particles  exhibit  random  orientation  and 
thus  prevent  light  transmission,  but  an  applied  electric  field  makes  the  orientation  orderly 
-and  permits  light  transmission  in  the  direction  of  the  field  (references  14,  15,  and  16). 
SCANNING  CIRCUITS  FOR  CATHODE-RAY  DEVICES.  The  electron  beam  of 
the  cathode-ray  tube  may  be  deflected  by  either  electric  or  magnetic  field.  Both  fields 
involve  the  storage  of  energy  in  the  deflection  space  within  the  tube  and  incidentally  in 
the  external  circuits. 

Wave  forms  for  a  deflecting  field  are  shown  in  Fig.  7.  Curve  A  shows  an  ideal  form. 
Curve  B  shows  a  departure  from  the  ideal  form  introduced  to  permit  a  finite  retrace  time 

and  thus  avoid  the  necessity  for  han- 
dling unreasonably  excessive  currents 
or  voltages  in  the  deflection  circuits 
during  the  retrace  intervals.  Curve  C 
shows  a  typical  practical  curve  in  which 
the  form  of  variation  is  modified  from 
that  of  curve  B  to  eliminate  the  discon- 
tinuities of  slope  exhibited  by  curve  B 
and  thus  reduce  the  number  of  harmon- 
ics which  must  be  faithfully  transmitted 
to  the  deflecting  circuit. 

For  faithful  reproduction  it  is  essen- 
tial that  the  scanning  wave -have  the 
same  form  in  both  the  pick-up  device 
and  the  reproducing  device.  It  has  not 


Time — ^- 

FIG.   7.     Scanning  Field  Wave  Forms.    A.  Ideal  Re- 
quirement.   B.  Modification  for  Finite  Retrace  Time. 
C,  Modification  for  Finite  Bandwidth. 


been  found  practicable  to  control  with  sufficient  accuracy  the  wave  form  of  the  deflecting 
field  during  the  retrace  interval.  It  is  essential,  therefore,  that  no  attempt  be  made  to 
transmit  picture  information  during  this  interval.  Means  are  customarily  provided  for 
preventing  the  appearance  of  the  reproducing  screen  from  being  affected  by  signal  potentials 
^hiring  the  retrace. 

SCANNING  CIRCUITS  FOR  ELECTROSTATIC  DEFLECTION.     Figure  8  shows  a 
typical  circuit  for  producing  electrostatic  deflection  voltages.     Oscillator  tube   V-l  is 


PICTUEE  DISPLAY  DEVICES 


20-09 


here  shown  as  a  blocking  oscillator.  Other  forms  such  as  multivibrators  and  thyratrons 
may  also  be  used.  The  oscillator  acts  periodically  to  discharge  capacitor  Ci.  The  ca- 
pacitor is  then  charged  through  R\  until  the  voltage  at  the  oscillator  anode  rises  to  a  point 
where,  in  conjunction  with  any  voltage  which  may  be  applied  on  the  grid,  oscillation  is 
again  produced,  thus  again  discharging  Ci.  The  voltage  variation  with  time  on  Ci  during 
the  trace  portion  of  the  period  is,  of  course,  exponential.  It  is  essential,  therefore,  that 
the  amplitude  at  this  point  be  kept  small  enough  to  preserve  a  good  approximation  to 


FIG.  8.    Electrostatic  Deflection  Circuit 

linearity.  The  amplifier  tube  Vz  and  phase  inverter  tube  Vz  amplify  the  voltage  appear- 
ing across  C\  to  the  required  level  and  provide  the  usually  necessary  outputs  of  opposite 
polarities.  In  choosing  the  values  of  components  care  must  be  taken  that  the  capacitors 
C2,  Cs,  C4,  and  Cs  do  not  introduce  excessive  phase  shift  for  the  fundamental  frequency 
component.  A  total  shift  of  3°  is  a  useful  upper  limit.  By  care  in  the  choice  of  com- 
ponent values  the  circuit  may  be  made  to  work  down  to  frequencies  much  lower  than  the 
normal  television  field  frequency  of  60  cycles;  it  may  also  be  used  readily  at  frequencies 
at  least  ten  times  higher  than  the  normal  line-scanning  frequency  of  15.7  kc. 

A  simpler  circuit  which  is  useful  for  scanning  frequencies  of  the  same  order  as  television 
line  frequencies  is  shown  in  Fig,  9.     In  this  circuit  capacitors  C\  and  Cz  are  charged  from. 


FIG.  9.    Electrostatic  Deflection  Circuit 


the  power  supply  during  the  trace  period  through  the  reactor  Li,  La.  While  this  charging 
current  is  oscillatory,  the  choice  of  a  resonant  frequency  less  than  one-tenth  the  scanning 
frequency  for  the  circuit  consisting  of  the  two  reactor  windings  and  two  capacitors  in 
series  results  in  the  use  of  only  the  linear  central  portion  of  the  oscillatory  cycle.  When 
the  voltage  appearing  across  Ci  and  Cz  is  great  enough,  tube  Vi  goes  into  oscillation  draw- 
ing a  heavy  current  and  discharging  the  pair  of  capacitors.  If  the  equivalent  series  re- 
sistance of  Vi  is  sufficiently  low,  this  discharge  will  be  oscillatory  and  will  last  for  one-half 
cycle  at  a  frequency  determined  by  the  apparent  inductance  of  the  primary  of  trans- 
former T  and  the  capacitance  of  capacitors  Ci  and  <72.  (Insufficient  leakage  inductance 
in  T  makes  the  retrace  time  unnecessarily  short  and  the  peak  current  and  dissipation  in 


20-10 


TELEVISION 


Vi  excessive.)  The  potential  difference  between  the  opposite  ends  of  Ci  and  C2  reverses 
during  this  discharge  and  rises  to  a  negative  value  which  may  be  a  considerable  fraction 
of  the  voltage  on  the  capacitors  before  Vi  began  to  conduct.  With  the  cessation  of  con- 
duction in  Vi,  Ci  and  C2  are  again  charged  through  I/i,  Z/2,  and  the  cycle  repeats. 

CIRCUITS  FOR  MAGNETIC  SCANNING.  In  the  magnetic  scanning  cycle  the  cur- 
rent in  the  coil  by  which  the  magnetic  field  is  produced  is  zero  at  the  midpoint  of  the 
scanning  trace.  During  the  latter  half  of  the  trace  the  current  is  built  up  under  control 

of  a  vacuum  tube  to  the  value  required  to  produce 
the  full  deflection.  The  flow  of  current  through 
the  vacuum  tube  is  then  interrupted,  and  the 
scanning  current  continues  to  flow  in  the  coil, 
charging  the  distributed  capacitance  which  is  in 
shunt  with  the  coil  until  the  current  vanishes. 
The  distributed  capacitance  is  now  charged  to  a 
high  potential;  it  discharges  back  into  the  coil, 
producing  a  current  of  opposite  polarity  to  that 
which  previously  flowed.  When  the  potential 
energy  stored  in  the  distributed  capacitance  has 
been  transferred  completely  back  to  the  coil,  the 
next  scanning-line  trace  starts;  the  flow  of  coil 
current  is  permitted  to  continue  through  a  vacuum 
tube  in  order  to  cause  the  current  to  decrease 
linearly  to  zero  value  at  the  midpoint  of  the  trace. 
In  this  cycle  energy  is  supplied  to  the  circuit  dur- 
ing the  latter  half  of  each  trace,  a  half-period  of 
free  oscillation  takes  place  during  the  retrace,  and 
the  energy  is  then  dissipated  during  the  first  half 
of  the  succeeding  trace.  These  relations  are 
shown  on  an  ideal  basis  in  Fig.  10.  It  is  necessary, 
of  course,  that  the  resonant  half-period  of  the  free 
oscillation  not  exceed  the  permissible  retrace  time. 
The  energy  required  in  the  magnetic  field  (with- 
in the  tube)  to  deflect  a  cathode-ray  tube  fully  is 
a  function  of  accelerating  voltage  and  deflection 
angle  primarily.  Typical  tubes  require  amounts 
of  energy  between  30  and  300  micro  joules.  These 
amounts  of  energy  must  be  provided  and  dissi- 
pated during  each  scanning  cycle.  Thus,  for  a 
type  10BP4  cathode-ray  tube  requiring  about  300 
microjoules  for  horizontal  deflection  and  a  hori- 
zontal deflection  frequency  of  15,750  the  power 
which  must  be  delivered  by  the  scanning  circuit 
within  the  tube  is  4.8  watts. 

A  typical  line-frequency  magnetic  scanning  cir- 
cuit is  shown  in  Fig.  11. 

A  step-down  transformer  is  interposed  between 
the  energy  supply  tube  V\  and  the  scanning  coil 
or  yoke  K,  to  decrease  the  effective  distributed 
capacitance  of  the  circuit,  thus  assisting  to  obtain 
suitably  short  retrace  time.  The  wave-form  dia- 
grams of  the  figure,  which  are  corrected  for  the 
effect  of  the  transformer  ratio,  show  the  current 
gradually  increasing  in  tube  V\  during  the  trace  and  terminating  abruptly  at  retrace.  Dur- 
ing this  same  period  the  current  in  tube  Vz  rises  abruptly  to  a  maximum  value  and  decreases 
smoothly  during  the  rest  of  the  trace.  The  analogy  between  these  two  current  wave 
forms  and  the  corresponding  forms  in  a  class  AB  amplifier  should  be  noted.  Tube  Vs 
provides  the  feedback  connection  for  self-oscillation. 

The  circuit  is  arranged  to  make  use  of  the  energy  returned  by  the  yoke  Y  during  the 
first  half  of  the  trace.  This  energy  is  stored  in  capacitor  Ci;  the  voltage  thus  developed 
across  Ci  is  used  to  augment  the  anode  supply  voltage  for  tube  Vi.  This  is  called  the 
"bootstrap"  connection. 

It  is  usually  necessary,  for  line-frequency  scanning,  to  use  a  relatively  low  inductance 
in  the  yoke.  The  current  is  therefore  relatively  high  and  may  reach  peak  values  of  0.1 
to  1:0  amp.  Relatively  large  direct  currents  (0.01  to  0.20  amp)  are  consequently  required 
to  correct  any  incidental  decentering  of  the  raster.  The  circuit  of  Fig.  11  is  therefore 


FIG.  10. 


Energy  Relations  in  Magnetic 
Scanning 


SYNCHRONIZING 


20-11 


designed  to  permit  the  full  B  supply  current  of  the  receiver  to  flow  through  the  centering 
control. 

Overall  efficiency  of  the  circuit  of  Fig.  11  may  reach  a  value  of  15  per  cent.  Delivery 
of  5  watts  to  the  magnetic  field  within  the  tube  thus  requires  at  least  30  watts  of  d-c  power 
(reference  80). 

The  power  required  for  the  relatively  slow  vertical  scanning  is  very  much  smaller  than 
that  for  horizontal  scanning.  A  typical  circuit  is  described  in  article  32  of  this  section. 
-JH 


Synchronizing 
e-f 


!FiG.  11.     Magnetic  Scanning  Circuit 

KEYSTONE  CORRECTION.  When  the  electron  beam  in  its  undeflected  position 
does  not  strike  its  target  at  a  right  angle  the  shape  of  the  raster  produced  by  uniform  de- 
flecting fields  will  be  a  keystone.  Circuit  arrangements  which  modulate  one  of  the  scan- 
ning generators  by  a  signal  derived  from  the  other  are  required  to  produce  a  rectangular 
raster  under  these  conditions.  Application  for  these  is  found  in  the  Iconoscope  type  of 
camera  tube  and  in  certain  forms  of  picture  tubes  (reference  81). 

6.  SYNCHRONIZING 

In  television  practice  the  picture  information  is  generated  in  an  orderly  sequence.  The 
picture  display  device  must  display  this  information  in  the  same  sequence  if  the  original 
picture  is  to  be  reproduced.  It  is  necessary,  therefore,  that  information  to  synchronize 
the  scanning  operations  of  the  display  device  be  furnished  with  the  picture  information 
and  that  this  information  be  subject  to  delays  in  transmission  identical  to  those  experienced 
by  the  picture  information.  Synchronizing  signals  are,  therefore,  included  with  the 
picture  signals.  ,  .  ,  . 

There  are  two  ways  in  which  scanning  devices  may  be  synchronized.  In  the  simpler  of 
these  the  synchronizing  signal  has  essentially  a  pulse  form  and  is  applied  to  the  scanning 
device  in  such  fashion  as  to  terminate  the  scanning  trace  and  initiate  the  retrace.  This 
action  takes  place  at  a  speed  limited  only  by  the  transient  response  of  the  scanning  oscil- 
lator itself  This  method  of  operation  has  the  advantage  of  simplicity  but  the  disad- 
vantage that  the  scanning  cycle  may  be  mistimed  and  the  picture  consequently  distorted 
by  either:  (1.)  a  noise  impulse  tripping  the  oscillator  prematurely;  (2)  loss  of  a  synchronizing 
pulse  due  to  a  temporary  blocking  of  the  signal  channel  by  noise,  or  (3)  the  combination 
of  random  noise  components  with  the  synchronizing  pulse  to  produce  random  phase  vari- 
ation of  the  leading  edge  of  the  pulse.  _  . 

The  alternative  synchronizing  method  is  to  apply  the  synchronizing  signal  and  a  signal 
derived  from  the  scanning  device  to  a  phase  comparison  circuit  whose  output  voltage 
controls  the  frequency  of  the  scanning  device.  If  the  synchronizing  pukes  are  uniformly 
spaced  and  the  scanning  device  is  itself  stable  in  frequency,  the  phase  control  may  be  made 
slow  acting,  thus  effectively  decreasing  the  band  width  of  the  synchronmng  signal  channel 
and  reducing  its  susceptibility  to  noise  interference  by  several  orders  of  magnitude  (refer- 
ence 17). 


20-12 


TELEVISION 


C 


a 


s 

•8 


o 

^ 


I 


I 


<  < 


<M 

QQ 


O 


CSJ 

O 


THE  VIDEO  SIGNAL 


20-13 


L 


Separate  synchronizing  signals  are  required  for  the  two  directions  of  scanning  With 
interlaced  scanning  it  is  essential  that  these  signals  be  readily  separable  one  from  the  other 
in  a  receiver.  The  form  of  composite  synchronizing  signal  which  has  been  adopted  to 
meet  these  requirements  is  shown  in  Fig.  12.  In  line  AI  of  Fig.  12  the  last  three  line- 
synchronizing  pulses  preceding  the  vertical  retrace  interval  appear  at  the  left,  followed 
by  six  equalizing  pulses,  six  field  synchronizing  or  "broad"  pulses,  and  six  more  equalizing 
pulses.  Transmission  of  normal  Line  pulses  is  then  resumed,  to  continue  until  the  next 
field  retrace  interval,  shown  at  A2.  The  traces  AI  and  A2  are  separated  in  time  by  exactly 
one  field  period;  in  consequence  of  the  use  of  "odd-line  interlace,"  the  line  pulses  in  AI  and 
A2  are  one-half  line  period  out  of  phase.  The  line-frequency  signals  may  be  separated 
from  the  composite  signal  by  differ- 
entiation, as  shown  at  BI,  B&;  the  f 1 It- 
arrows  here  represent  the  times  at 
which  the  line  oscillator  should 
"fire";  at  each  of  these,  a  synchro- 
nizing signal  is  provided.  The  field 
frequency  signals  may  be  separated 
by  integration,  with  the  results 
shown  at  Ci,  Cj.  It  is  a  characteris- 
tic of  an  integrating  circuit  that  it 
"remembers."  For  this  reason  the 
interval  immediately  preceding  the 
vertical  synchronizing  signal  con- 
tains horizontal  synchronizing  sig- 
nals at  twice  the  normal  repetition 
rate.  By  this  means  the  time  in- 
tervals immediately  preceding  the 
vertical  synchronizing  pulses  in  the 
two  fields  are  made  identical.  The 
line  synchronizing  pulses  are  reduced 
to  half  their  normal  duration  during 
this  period  so  that  their  integrated 
value  will  be  no  greater  than  that 
of  line  pulses  of  normal  duration 
and  normal  repetition  rate.  These 
equalizing  pulses  also  appear  for  a 


0         ^ 

r  i   = 

+B 

>           '  o 

-a 

Differentiating  C 

¥• 

ircuit  for  Line;    RC  =  2.  usec. 

Integrating  Circuit  tor  Field;  R=!00k  ohms, 
C«  0.005  jif. 

FIG.  13.     Synchronizing  Signal  Selection 


short  interval  following  the  vertical  synchronizing  signal  to  insure  that  during  the  entire 
interval  in  which  the  vertical  scanning  device  is  sensitive  to  synchronizing  signals  those 
signals  will  be  alike  in  both  fields. 

The  diagrams  of  Fig,  13  illustrate  differentiating  and  integrating  circuits  to  perform 
the  separations  shown,  in  Fig.  12. 


7.  THE  VIDEO  SIGNAL 

The  video  signal  is  generated  by  a  pick-up  tube  as  described  in  Section  15.  The  output 
of  this  device  usually  requires  amplification  to  raise  it  to  usable  level  and  may,  in  addition, 
require  processing  to  remove  from  the  signal  certain  spurious  components  which  are  not 
properly  a  part  of  the  signal.  Its  direct  component  must  either  be  transmitted  faith- 
fully with  the  same  gain  as  other  components  or  be  reinserted  by  either  manual  or  auto- 
matic means  after  amplification  has  taken  place. 

Figure  14  shows  a  test  pattern  used  for  testing  television  systems  and  the  oscillogram 
of  a  single  scanning  line  of  that  pattern.  The  oscillogram  was  taken  at  a  point  in  the 
transmission  system  where  all  spurious  components  had  been  eliminated.  The  corre- 
spondence between  elements  of  the  picture  and  elements  of  the  oscillogram  is  shown. 

TRANSMISSION  OF  THE  D-C  COMPONENT.  It  is  theoretically  possible  to  trans- 
mit and  amplify  the  d-c  component  along  with  the  other  components  of  the  video  signal. 
In  practice,  however,  this  is  frequently  found  to  be  inconvenient.  A  satisfactory  alter- 
native, known  as  "d-c  reinsertion,"  may  be  followed  once  the  black  level  of  the  signal  has 
been  established.  In  this  alternative  practice  the  recurring  black  intervals  are  used  to 
provide  an  a-c  carrier  of  the  d-c  component.  This  practice  is  illustrated  in  its  simplest 
form  in  the  two-stage  amplifier  of  Fig.  15.  The  blocking  capacitor  Ci  prevents  transmis- 
sion of  the  d-c  component  from  the  anode  circuit  of  Vi  to  the  grid  of  "V&  The  video  signal 
is  applied  to  the  grid  of  Yi  with  positive  polarity,  that  is,  with  an  increase  in  object  bright- 
ness represented  by  a  change  of  signal  potential  in  the  positive  direction.  The  black  level, 


20-14 


TELEVISION 


therefore,  represents  the  most  negative  portion  of  the  signal.  The  signal  polarity  is  re- 
versed in  the  anode  circuit  of  Vi  so  that  at  this  point  the  black  level  is  the  most  positive 
portion  of  the  signal.  The  diode  Vz  is  connected  to  conduct  on  the  positive  portion  of 


FIG.  14.     Television  Test  Pattern  and  Sample  Line  Wave  Form.     Test  pattern  (copyrighted  by  Radio 
Corp.  of  America).    Wave  form  of  single  line  as  shown. 

the  signal  reaching  it.  (The  grid-to-cathode  conductance  of  Vz  shows  a  similar  character- 
istic.) Current  flowing  through  the  diode  in  response  to  the  black  intervals  of  the  signal 
establishes  such  a  charge  on  Ci  as  to  reduce  the  flow  of  current  through  the  diode  to  a  value 
just  sufficient  to  make  up  for  the  leakage  from  the  condenser  through  Rit  If  the  effective 
resistance  of  the  charging  path  (including  the  series  resistance  of  the  diode  and  the  re- 
sistance of  Rz)  is  small,  the  amount  by  which  the  diode  anode  is  positive  with  respect  to  its 


FIG.  15.     D-c  Reinsertion  Circuit 


cathode,  during  the  black  intervals,  will  be  a  negligible  fraction  of  the  total  signal  poten- 
tial. The  black  intervals  of  the  signal  will,  therefore,  be  held  at  substantially  the  poten- 
tial of  the  diode  cathode,  and  the  gray  and  white  portions  will  extend  negatively  from  this 
potential  in  their  appropriate  amounts. 


THE    VIDEO   SIGNAL 


20-15 


2g 


§     i     §     i 

*+••!•  + 


-^        m-1 


T       i      T 


>  2 


1 

(apouv) 


I 


s 


8 

CJ 

•f- 


(apouy) 


0> 


20-16 


TELEVISION 


Signal 
Input 


Figure  16  shows  the  performance  of  the  circuit  of  Fig.  15  with  two  different  signals. 
Signal  A  has  a  little  white,  but  mostly  dark  gray  and  black,  while  signal  B  is  all  white 
except  for  a  brief  black  interval.  The  effectiveness  of  the  reinsertion  action  is  shown  by 
the  substantial  agreement  in  black  levels  for  both  signals,  at  the  grid  and  also  at  the  anode 
of  72.  The  curves  shown  for  the  performance  at  Vz  when  the  diode  is  replaced  by  a  fixed 

bias  show  the  much  greater  range  of  voltages 
which  tube  Vi  must  handle  without  distortion, 
if  d-c  reinsertion  is  not  practiced* 

Because  of  the  high  impedance  of  diodes  at 
the  low  currents  they  are  called  upon  to  handle 
in  the  circuit  of  Fig.  15,  the  accuracy  with 
which  the  black  level  is  maintained  is  some- 
what imperfect.  When  more  accurate  per- 
formance is  required,  a  more  elaborate 
arrangement  known  as  a  clamp  circuit  may  be 
used  in  place  of  the  diode  Vz  and  resistor  RI  of 
Fig.  15.  In  a  clamp  circuit,  an  external  source 
applies  power  to  a  network  of  diodes,  producing 
considerable  currents  and  low  diode  imped- 
ances. The  network  is  balanced  so  that  any 
deviation  of  the  signal  potential  from  the 
potential  of  a  reference  point  during  the  flow 
of  current  from  the  external  source  is  corrected 
by  a  small  unbalance  in  the  currents  in  the 
branches  of  the  circuit.  Figure  17  shows  one 
form  of  clamp  circuit.  Clamp  circuits  differ 
from  the  circuit  Fig.  15  in  that  (a)  they  are 


Reference 
'  Potential 


JUL' 


^Energizing 
Signal 


FIG.  17.    Clamp  Circuit 


capable  of  conduction  in  both  directions;  (6)  in  consequence  of  this  they  must  be  energized 
•nly  during  appropriate  time  intervals.    See  also  reference  27  and  84. 


8.  THE  COMPOSITE  SIGNAL 

la  the,  design  of  television  systems  provision  must  be  made  for  the  transmission  of  these 
four  signals:  (a)  video  signal;  (6)  horizontal  synchronizing  signal;  (c)  vertical  synchronizing 
signal;  and  (d)  sound  signal. 

The  system  may  be  designed  to  transmit  all  four  of  these  from  separate  transmitters. 
Alternatively,  two  or  more  may  be  combined  and  transmitted  by  a  single  transmitter. 
The  combination  of  the  picture  signal  and  the  two  synchronizing  signals  in  a  single  trans- 
mission has  been  recognized  as  particularly  suitable,  since  it  simplifies  both  receiving  and 
transmitting  apparatus  and  also  removes  some  sources  of  non-uniform  transmission  delay 
between  these  components. 

The  construction  of  a  composite  signal  containing  these  three  individual  signals  requires 
the  synchronizing  and  picture  signals  to  occupy  different  ranges  of  amplitude,  since  these 
two  classes  of  signals  cannot  be  distinguished  from  one  another  by  a  frequency  separation. 
They  must  also  occupy  different  time  intervals.  These  requirements  are  satisfied  by 
assigning  a  range  of  potentials  beyond  black  (and,  therefore,  called  infra-black)  to  the 
synchronizing  signals  and  by  inserting  synchronizing  signals  in  the  time  intervals  provided 
for  scanning  retrace.  Figure  18  shows  at  3  an  oscillogram  of  two  lines  of  a  composite 
signal  showing  line  synchronizing  pulses  properly  located  in  the  retrace  intervals.  The 
position  of  the  leading  edge  of  the  pulse  in  the  retrace  interval  is  set  a  short  time  after  the 
beginning  of  the  interval  so  that  even  receiver  circuits  of  somewhat  restricted  band  width 
(and  hence  slow  transient  response)  will  have  time  to  reach  black  level  before  the  syn- 
chronizing pulse  begins,  regardless  of  whether  the  picture  edge  is  white  or  black.  Failure 
to  provide  this  interval  results  in  phase  modulation  of  the  scanning  by  the  picture  content. 
This  interval  (sometimes  called  the  "front  porch")  is  made  no  greater  than  required  by  the 
foregoing  consideration,  since  any  extra  waiting  at  this  point  is  at  the  expense  of  either 
decreased  time  available  for  scanning  retrace  or  decreased  time  available  for  picture. 

The  placement  of  the  field  signal  conforms  to  the  practice  already  described  for  the  line 
signal.  The  field  signal  is  located  in  the  same  region  of  amplitude  as  the  line  signal  and  is 
placed  to  occur  during  the  vertical  retrace  region. 

The  portion  of  the  transmission  amplitude  range  not  occupied  by  synchronizing  signals 
is  reserved  for  the  picture  information. 


THE  RADIO-FBEQTJENCY  SIGNAL 


20-17 


Picture-^]  TL   T 
Hor.  blanking— -1}*- 


w  Vertical 
blanking  0.05  7°'03V 
4— Bottom  of     ~ 

picture  Tim- 

I 


« — -Top  of  picture 


nftnnnni  1 1 1 1 1  i..i  i  n 


Horizontal  dimensions  not  to  scaJe 


Black 
3         level 

Detail 
between 
3<3  In  2. 

White  level- 
Zero  carrier 


G)  0.004  H_» 
max. 


Detail 
between 
4-4  in  2. 


Black  level-*- 
(Z)0.04  H— > 
See  note  6 


<-       (u)  0.004  H-* 
max, 
-v|L(&)  0.004  H 

(p)  0.004  H., 
max. 

~* 

^(5)0.004  H 
max. 

^-Equalizing 
pulse 

Vertical 
sync,  pulse 

f,   9/10  of 
max.  sync. 

J,     1/10  of 

iW)o.5Hij 

f       .,!-!,  h 

—jj-max.  sync. 
<Ar)  0.07  H±  0.01  H* 

(s)0.004  H 


(«)0,Q04  H 


NOTE: 

1.  H=  Time  from  start  of  one  line  to  start  of  next  line. 

2.  V  —Time  from  start  of  one  field  to  start  of  next  field. 

3.  Leading  and  trailing  edges  or  vertical  blanking 
should  be  complete  In  less  than  0*1  H. 

4.  Leading  and  trailing  slopes  of  horizontal  blanking 
must  be  steep  enough  to  preserve  mln.  and  max. 
values  of  (as-fl/)  and  (£)  under  all  conditions  of 
picture  content* 

*5.  Dimensions  marked  with  an  asterisk  Indicate  that 
tolerances  given  are  permitted  only  for  long  time 
variations,  and  not  for  successive  cycles. 

6.  Equalizing  pulse  area  shell  be  between  0.45  and 
0.5  ot  the  area  of  a  horizontal  sync,  pulse. 

7.  Refer  to  text  for  further  explanations  a.nd 
tolerances. 

FIG.  18.     FCC  Standard  Television  Synchronizing  Wave  Form 


min. 


9.  THE  RADIO-FREQUENCY  SIGNAL 

The  composite  signal  of  Fig.  18  may  be  applied  to  an  r-f  carrier  as  either  amplitude, 
phase,  or  frequency  modulation.  In  television  broadcasting,  multipath  transmission  is 
frequently  observed;  picture  distortions  caused  by  multipath  transmission  when  phase  or 
frequency  modulation  is  used  are  so  serious  that  these  methods  of  modulation  have  not 
seemed  practicable.  Television  broadcasting,  therefore,  makes  use  of  amplitude  modula- 
tion. 


20-18 


TELEVISION 


.2 

£ 


O 

O. 


1 


.i 

"5 
CD 


THE  K4DIO-FREQUENCY  SIGNAL 


20-19 


POLARITY  OF  MODULATION.  Polarity  of  modulation  may  be  either  positive  (that 
is,  with  an  increase  of  image  brightness  represented  by  an  increase  of  radiated  signal)  or 
negative.  Figure  19  illustrates  these  two  forms,  showing  individual  scanning  line  signals 
for  three  distributions  of  picture  content.  A  positive  modulation  polarity  signal  includes 
at  all  times  the  synchronizing  level  (zero  carrier)  and  the  black  level.  It  does  not  indicate 
the  level  of  peak  white  unless  elements  of  this  intensity  are  present  in  the  picture.  Neg- 
ative modulation  polarity,  on  the  other  hand,  includes  at  all  times  the  synchronizing  level 
(maximum  carrier  intensity),  the  black  level,  and  peak  white  (zero  carrier). 

Automatic  gain  Control  circuits  for  receivers  require  the  presence  in  the  received  signal 
of  some  characteristic  which  is  independent  of  modulation.  In  sound  transmissions,  the 
average  value  of  the  carrier  has  the  required  characteristic,  but  in  television  signals,  the 
average  value  is  dependent  on  average  picture  brightness.  White  level,  black  level,  or 
synchronizing  level  must  be  used  instead.  Preferably,  the  peaks  of  the  signal  envelope 
should  be  used,  so  that  a  simple  peak  detector  may  serve  as  the  source  of  automatic  gain 
control  information.  It  is  found,  therefore,  that  negative  modulation  polarity  simplifies 
very  much  the  provision  of  automatic  gain  control  in  receivers. 

The  effects  of  impulse  noise  interference  on  signals  of  the  two  polarities  are  quite  dif- 
ferent. With  positive  modulation  impulse  noise  usually  produces  bright  spots  in  the  re- 
produced picture  and  has  little  effect  on  synchronizing  signals.  With  negative  modula- 
tion impulse  noise  produces  primarily  black  spots  on  the  picture  (which  are  on  the  whole 
less  disturbing  than  the  bright  spots  produced  with  positive  modulation)  but  has  a  greater 
tendency  to  interfere  with  synchronizing  signals.  Since  it  is  found  possible  to  minimi&e 
the  effect  of  impulse  noise  on  synchronizing  sufficiently  by  careful  circuit  design  in  the  re- 
ceiver and  since  automatic  gain  control  is  believed  desirable,  American  standards  for  tele- 
vision have  chosen  negative  modulation  polarity. 

BAND  WIDTH.  As  was  shown  in  article  3,  the  desired  band  width  of  television  video 
signals  exceeds  4  megacycles.  The  application  of  this  signal  as  amplitude  modulation  to 
a  carrier  produces  a  signal  having  a  total  spectrum  width  which  exceeds  8  megacycles. 
Since  radio  channels  are  not  available  in  unlimited  quantities  and  since  also  the  cost  of 
amplifiers  is  increased  as  their  required  band  width  increases,  television  broadcast  practice 
is  based  on  vestigial  sideband  transmission.  Curves  A  and  B  of  Fig.  20  show  the  radio- 


1.0 


0.5 


0.0 


B 


-5 


-3-2-1  O  I  2 

Frequency  Relative  to  Picture  Carrier 


01  23456 

Frequency  Relative  to  Assigned   Channel  Boundaries 

FIG.   20      Radio-frequency  Amplitude   Characteristics.     A.    Double    Side-band    Transmission.     B. 
Vestigial   Side-band   Transmission   Overall   Characteristic.     C.   Vestigial    Side-band    Transmission; 

Transmitter  Only. 

frequency  amplitude  characteristics  for  double  sideband  transmission  and  for  vestigial 
sideband  transmission  (references  19  to  24) . 

The  overall  transmission  characteristic  for  vestigial  sideband  transmission  requires  that 
the  signal  originally  produced  with  a  carrier  and  symmetrical  sidebands  must  be  atten- 


20-20  TELEVISION 

uated  selectively.  The  practice  which  has  been  standardized  Is  to  provide  a  receiver 
characteristic  having  the  same  form  as  the  desired  system  overall  characteristic.  The 
corresponding  transmitter  characteristic  must,  therefore,  exhibit  negligible  attenuation 
at  all  frequencies  which  are  effectively  transmitted  by  the  receiver.  It  has  therefore  been 
standardized  as  shown  in  Fig.  20  as  curve  C. 

SOUND  TRANSMISSION.  The  sound  accompanying  a  television  picture  is  trans- 
mitted on  a  separate  carrier  whose  frequency  is  located,  with  respect  to  the  picture  carrier 
and  its  sidebands,  as  shown  in  Fig.  20.  The  sound  carrier  is  frequency  modulated  with 
maximum  deviation  of  25  kilocycles.  The  pre-emphasis  practice  standardized  for  fre- 
quency-modulated sound  broadcasting  is  also  used  for  television  sound. 

FREQUENCY  ALLOCATION.  Because  of  the  wide  frequency  channels  required  for 
television,  allocation  of  channels  below  50  megacycles  would  interfere  with  so  many  exist- 
ing services  as  to  be  impracticable.  Television  allocations  for  commercial  use,  therefore, 
lie  in  the  range  between  54  and  216  megacycles,  as  shown  in  the  table.  Allocations  at 
higher  frequencies  have  been  made  for  experimental  and  relay  use. 

NOMINAL  PICTUEB  SOUND 

CHANNEL     BOUNDARIES  CARRIER  CARRIER 

2  54-  60  55.25  59.75 

3  60-  66  61.25  65.75 

4  66-  72  67.25  71.75 

5  76-  82  77.25  81.75 

6  82-  88  83.25  87.75 

7  174-180        175.25        179.75 

8  180-186        181.25        185.75 

9  186-192        187.25        191.75 

10  192-198  193.25  197.75 

11  198-204  199.25  203.75 

12  204-210  205.25  209.75 

13  210-216  211.25  215.75 

POLARIZATION  OF  RADIATED  SIGNAL.  A  simple  horizontally  polarized  dipole 
antenna  has  a  horizontal  directive  pattern  which  is  sometimes  useful  in  minimizing  ef- 
fects- of  multiple  transmission  paths.  Since  in  other  respects  there  is  little  net  advantage 
either  way  between  horizontal  and  vertical  polarization,  horizontal  polarization  has  been 
standardized. 

10.  STANDARDS 

The  Federal  Communications  Commission  has  established  the  following  Standards  of 
Good  Engineering  Practice  for  television  broadcasting: 

1.  The  width  of  the  television  broadcast  channel  shall  be  6  megacycles  per  second. 

2.  The  visual  carrier  shall  be  located  4.5  megacycles  lower  in  frequency  than  the  aural  center  fre- 
quency. 

3.  The  aural  center  frequency  shall  be  located  0.25  megacycle  lower  than  the  upper  frequency  limit 
of  the  channel. 

4.  The  visual  transmission  amplitude  characteristic  shall  be  as  shown  in  Appendix  II  [curve  C  of 
Fig.  20]. 

5.  The  number  of  scanning  lines  per  frame  period  shall  be  525,  interlaced  2:1. 

6.  The  frame  frequency  shall  be  30  per  second,  and  the  field  frequency  shall  be  60  per  second. 

7.  The  aspect  ratio  of  the  transmitted  television  picture  shall  be  4  units  horizontally  to  3  units  verti- 
cally. 

8.  During  active  scanning  intervals,  the  scene  shall  be  scanned  from  left  to  right  horizontally  and  from 
top  to  bottom  vertically,  at  uniform  velocities. 

9.  A  carrier  shall  be  modulated  within  a  single  television  channel  for  both  picture  and  synchronizing 
signals,  the  two  signals  comprising  different  modulation  ranges  in  amplitude  (see  Appendices  I  and  II) 
[Figs.  18  and  20]. 

10.  A  decrease  in  initial  light  intensity  shall  cause  an  increase  in  radiated  power  (negative  trans- 
mission). 

11.  The  black  level  shall  be  represented  by  a  definite  carrier  level,  independent  of  light  and  shade  in 
the  picture. 

12.  The  pedestal  level  (normal  black  level)  shall  be  transmitted  at  75  per  cent  (with  a  tolerance  of 
plus  or  mintis  2.5  per  cent)  of  the  peak  carrier  amplitude. 

13.  The  maximum  white  level  shall  be  15  per  cent  or  less  of  the  peak  carrier  amplitude. 

14.  The  signals  radiated  shall  have  horizontal  polarization. 

15*  A  radiated  power  of  the  aural  transmitter  not  less  than  50  per  cent  or  more  than  150  per  cent  of 
the  peak  radiated  power  of  the  video  transmitter  shall  be  employed. 


LENS  APERTURE  REQUIRED  20-21 

^  16.*  Variation  of  Output.  The  peak-to-peate  variation  of  transmitter  output  within  one  frame  of 
video  signal  due  to  all  causes,  including  hum,  noise,  and  low-frequency  response,  measured  at  both 
synchronizing  peak  and  pedestal  level,  shall  not  exceed  5  per  cent  of  the  average  synchronizing  peak 
signal  amplitude. 

17.*  Black  Level.  ^  The  black  level  should  be  made  as  nearly  equal  to  the  pedestal  level  as  the  state 
of  the  art  will  permit.  If  they  are  made  essentially  equal,  satisfactory  operation  -will  result  and  im- 
proved techniques  will  later  lead  to  the  establishmejxb  of  the  tolerance  if  necessary, 

18.*  Brightness  Characteristics.  The  transmitter  output  shall  vary  in  substantially  inverse  loga- 
rithmic relation  to  the  brightness  of  the  subject.  No  tolerances  are  set  at  this  time, 

See  also  reference  28  for  an  extended  discussion  of  standards. 


TELEVISION  BROADCASTING 

By  T.  J.  Buzalski,  A- I,,  Hammcrschjnidt,  apd  F,  J.  Somers 

A  modern  television  broadcasting  plant  provides  facilities  for  the  pick-up  and  broad- 
cast of  entertainment,  news,  and  cultural  and  educational  subject  matter  in  both  sight 
and  sound.  The  purpose  of  such  a  plant  is  to  provide  an  adequate  and  satisfactory  public 
service,  and  this  requires  a  flexible  and  well-coordinated  installation.  A  functional  sub- 
division of  equipment  and  facilities  is  the  following: 

(a)  Studio  and  control  facilities  (picture  and  sound) . 

(6)  Field  pick-up  and  relay  facilities  (picture  and  sound). 

(c)  Visual  and  aural  broadcast  transmitters. 

Typical  studio  and  control  facilities  consist  of  one  or  more  live-talent  studios,  a  film 
pick-up  studio,  a  video  effects  studio,  one  or  more  announcers'  booths,  and  a  master  con- 
trol room  having  switching  and  monitoring  facilities  for  feeding  the  various  studio  outputs 
or  remote  pick-up  outputs  to  the  transmitter,  as  required.  The  master  timing  or  syn- 
chronizing generator,  various  picture  Hue  amplifiers,  power  supply  rectifiers,  and  other 
equipment  common  to  the  studio  facilities  system  are  usually  grouped  in  a  main  equip- 
ment room  for  maximum  efficiency  and  ease  of  maintenance. 

Field  pick-up  facilities  include  portable  television  cameras  with  their  associated  control, 
monitoring,  and  synchronizing  equipment  and  portable  sound  equipment.  Either  radio 
relay  circuits,  coaxial  cables,  or  equalized  telephone  lines  are  used  to  transmit  the  picture 
signals  back  to  the  master  control  room  of  the  broadcast  station  proper.  The  sound  por- 
tion of  the  program  is  generally  transmitted  back  by  wire  line,  though  radio  circuits  are 
used  where  wire  facilities  are  not  available.  Field  pick-ups  also  encompass  the  use  of 
mobile  equipment  where  the  television  cameras,  along  with  their  synchronizing,  control, 
and  monitoring  equipment,  are  mounted  in  a  moving  boat,  aircraft,  or  other  means  of 
locomotion. 

The  need  for  maximum  height  of  the  transmitting  antenna  to  provide  line-of-sight  re- 
ception for  as  many  receivers  as  possible  usually  requires  that  the  visual  and  aural  trans- 
mitters be  located  remote  from  the  television  studios.  The  visual  lirik  between  the  master 
control  switching  point  and  the  transmitter  may  be  a  radio  relay  circuit,  a  coaxial  cable, 
or  an  equalized  telephone  line.  The  aural  link  between  the  master  control  point  and  the 
transmitter  is  usually  a  wire  line. 

In  order  to  coordinate  operations  and  to  assure  program  continuity,  the  television  plant 
must  be  provided  with  an  adequate  and  flexible  intercommunication  and  order  wire  sys- 
tem separate  and  apart  from  the  sound  program  pick-up,  control,  and  transmission  equip- 
ment. 

1L  LENS  APERTURE  REQUIRED 

The  lens  speed  required  for  a  given  camera  pick-up  tube  and  scene  illumination  is  best 
determined  by  experiment  under  operating  conditions  or  from  data  supplied  by  the  tube 
manufacturer.  If  it  is  desired  to  compute  the  lens  speed  which  will  provide  a  sufficiently 
bright  image  to  meet  the  requirements  of  a  given  pick-up  tube,  the  following  formulas 
{see  reference  33)  are  applicable: 

F  (focal  length  of  lens)  =  —  cot  ^  inches  (1) 

aperture) 


*  These  items  are  subject  to  change  but  are  considered.the  best  practice  under  the  present  s*ate  of 
the  art.    They  will  not  be  enforced  pending  a  further  determination  thereof. 


20-22 


TELEVISION 


where  W  =  the  width  in  inches  of  the  pick-up  tube  sensitive  surface. 
a  =  the  desired  horizontal  angle  of  view  in  degrees, 
s  =  the  pick-up  tube  sensitivity  in  signal  microamperes  per  lumen. 
B  —  the  surface  brightness  of  the  scene  in  candles  per  square  foot. 
T  —  the  light  transmission  factor  of  the  lens  (usually  between  40  and  60  per  cent) . 
N  —  the  required  peak  picture  signal-to-rms  noise  ratio. 
7n  =  the  equivalent  rms  noise  current  (amperes)  at  the  input  of  the  amplifier  used 

with  the  pick-up  tube. 

The  noise  generated  in  pick-up  tubes  without  electron  multipliers  is  small  compared 
with  that  originating  in  the  first  tube  of  the  video-preamplifier.  Noise  currents  generated 
in  the  first  video  amplifier  stage  result  from  two  significant  components :  thermal  agitation 
noise  in  the  input  circuit  and  current  fluctuations  in  the  plate  circuit  of  the  first  video 
amplifier  tube.  An  expression  combining  these  components  for  the  computation  of  the 
equivalent  rms  noise  current  ( Jn)  follows  (see  reference  33) : 

^  (3) 


where  In  =  equivalent  rms  noise  current  in  amperes. 

k  =  Boltzmann  constant  (1.37  X  10~23  joule  per  °K). 
T  «  absolute  temperature  (300°  K). 
fm  —  pass  band  of  amplifier  in  cycles  per  second. 
R  —  input  resistor  of  amplifier  in  ohms. 
Rt  =  equivalent  grid  resistance  of  input  tube  for  noise  in  ohms  at  300°  K  (for  first 

video  amplifier  tube). 

C  —  total  shunt  capacity  (pick-up  tube,  stray  capacity,  and  video  amplifier  in- 
put capacity). 

In  a  non-storage  pick-up  tube  where  the  photoelectrons  are  amplified  by  an  electron 
multiplier  incorporated  in  the  pick-up  tube,  the  predominant  noise  originates  at  the 
photo  cathode.  (See  reference  34.)  The  equivalent  rms  noise  cm-rent  in  this  case  is  given 

by  the  following  expression:  

In  —  ^/2eiofm  amperes  (4) 

where  io  =  current  for  one  picture  element. 

e  -  electron  charge  =  1.59  X  10~19  coulomb. 
fm  —  pass  band  of  equipment  (cycles  per  second) . 

It  should  be  pointed  out  that  the  sensitivity  s  of  a  pick-up  tube  in  signal  microamperes 
per  lumen  is  not  necessarily  a  constant.  In  some  types  of  tubes  «  is  a  function  of  the  in- 
cident light.  This  is  illustrated  by  the  curve  of  Fig.  1,  which  shows  s  for  a  sample  Icono- 


.150 
£.125 


•5.010 


o.075 


&.050 

CO 

.025 


0  J.  .2  .3  .4  .5  6  .7  .8  .9  ,10 

Light  on  mosaic,  lumens 

FIG.  1.     Signal  Output  vs   Illumination  Characteristic  of  a  Typical  Iconoscope 

scope  (reference  56).  Another  type  of  tube  having  a  non-linear  sensitivity  curve  is  the 
Image  Orthicon  (reference  30) .  The  signal  output  vs.  illumination  curves  of  the  Orthicon 
and  the  Image  Dissector  are  essentially  straight  lines. 


12.  STUDIO  CAMERA  DESIGN 

Aside  from  the  pick-up  lens,  view  finder  (references  31  and  32),  and  other  optical  re- 
quirements, the  electrical  and  mechanical  design  of  the  studio  camera  must  be  given  care- 
ful consideration.  Experience  has  shown  that  the  following  principles  should  be  followed 
in  the  design: 


STUDIO  CAMERA  DESIGN  20-23 

(a)  In  order  to  keep  the  camera  as  small,  light,  and  mobile  as  possible,  the  number  of 
tubes  and  components  housed  in  the  camera  proper  should  be  kept  to  the  mim'rnttm.  As 
much  auxiliary  equipment  as  possible  should  be  mounted  on  permanent  racks  in  or  near 
the  control  room. 

(&)  All  electrical  camera  controls  and  adjustments  which  may  require  attention  during 
a  broadcast  should  be  located  at  or  within  easy  reach  of  the  control-room  console. 

(c)  The  mechanical  layout  of  the  camera  should  facilitate  rapid  servicing  in  case  of 
failure.  Such  features  as  replaceable  plug-in  type  video  preamplifiers  will  simplify  main- 
tenance of  the  camera. 

A  variety  of  different  types  of  camera  tubes  have  been  used  for  television  studio  pick-up. 
Before  1946  the  Iconoscope  camera  found  the  widest  acceptance  among  television  broad- 
casters. Subsequently  the  Image  Orthicon  (reference  30)  with  its  greater  light  sensitivity 
has  been  found  to  have  practical  advantages  for  studio  use.  Since  space  does  not  permit 
a  detailed  discussion  of  all  camera  types,  the  Iconscope  has  been  chosen  in  the  following 
example  since  it  embodies  many  features  that  are  common  to  present  television  cameras. 

In  a  typical  Iconoscope  studio  camera  design,  the  following  items  are  included  in  the 
camera  housing  proper: 

(a)  Video  preamplifier  (five  tubes)  including  a  high-peaker  stage  and  a  cathode  follower 
to  feed  a  75-ohm  coaxial  line. 

(6)  Beam  blanking  amplifier  (double  triode). 

(c)  Horizontal  and  vertical  deflection  coils  in  a  yoke  assembly  arranged  to  be  fed  from 
an  external  source  via  coaxial  cables. 

(d)  Bias  lighting  arrangement  for  improving  the  collection  of  secondary  electrons  from 
the  mosaic. 

(e)  Filament  transformer  (110-volt  alternating  current  to  6.3-volt  alternating  current 
60  cycles). 

This  particular  design  of  camera  used  an  optical  view  finder  and  was  mounted  on  a 
movable  pedestal  equipped  with  a  pushbutton-controlled  motor-driven  elevating  arrange- 
ment, a  suitable  tilting  and  panning  head  being  provided  for  mounting  the  camera. 

A  typical  flexible  cable  for  connecting  the  camera  to  the  rack  equipment  has  conductors 
and  insulation  as  shown  in  Table  1.  As  indicated  in  the  table,  this  cable  has  been  designed 
as  an  all-purpose  cable  which  can  be  used  with  a  variety  of  camera  and  pick-up  tube  types. 
Conductors  26  and  27  are  coaxial  cables,  conductor  group  28,  29  is  a  shielded  balanced 
twisted  pair  video  cable.  Conductors  3  and  4  are  used  to  feed  high  voltage  to  the  pick-up 
tube.  Video  cable  shields  should  be  run  separately  from  the  common  power  supply  cir- 
cuit ground  to  avoid  possible  interference  pick-up  due  to  common  ground  return  imped- 
ance. 

i  For  camera  cable  lengths  of  100  to  150  ft  maximum,  it  is  practical  to  locate  the  deflection 
wave  generators  in  the  control-room  racks,  provided  sufficiently  low-impedance  deflection 
coils  are  used  in  the  camera  -deflection  yoke.  For  longer  cable  lengths,  it  is  advisable  to 
locate  the  deflection  amplifiers  in  the  camera  proper  or  in  the  camera  pedestal,  feeding  the 
horizontal  and  vertical  driving  pulses  or  sawtooth  waves  out  to  the  camera  from  the  racks 
via  terminated  75-ohm  coaxial  cables  or  terminated  balanced  pair  twinax  cables. 

Some  types  of  camera  tubes  are  easily  damaged  if  the  deflection  circuits  fail  during 
operation.  This  is  true  of  Iconoscopes.  Orthicons  may  also  be  damaged  if  the  mosaic  is 
illuminated  when  the  deflection  failure  occurs.  It  is,  therefore,  necessary  to  provide  pro- 
tective devices  to  bias  off  the  scanning  beam  automatically  when  deflection  fails.  Relays 
actuated  by  rectified  deflection  currents  have  proved  satisfactory  for  this  purpose. 

The  video  output  of  the  camera  may  be  fed  to  the  control  room  from  the  camera  either 
by  a  coaxial  line  or  by  a  balanced  twisted  pair  transmission  line  with  an  external  grounded 
shield.  The  latter  is  preferred  to  give  better  discrimination  against  hum  pick-up  in  long 
cables. 

The  camera  preamplifier  design  usually  provides  for  a  peak-to-peak  video  output  from 
the  camera  of  between  0.1  and  1.0  volt.  The  higher  levels  are  preferred  to  provide  dis- 
crimination against  possible  interference  pick-up  in  the  cable. 

VIDEO  PREAMPLIFIER  DESIGN.  The  scene  brightness,  the  speed  of  the  lens,  the 
sensitivity  of  the  pick-up  tube,  and  the  value  of  pick-up  tube  signal  load  resistor  being 
known,  the  low-frequency  gain  required  of  the  preamplifier  to  bring  the  signal  up  to  the 
desired  level  to  feed  the  75-ohm  cable  can  readily  be  calculated.  (By  low-frequency  gain 
is  meant  the  gain  for  video  frequencies  below  50  or  100  kc.)  The  high-frequency  gain  re- 
quired of  the  amplifier,  assuming  that  the  general  practice  of  feeding  a  "flat"  signal  to  the 
line  is  followed,  is  dependent  on  the  frequency  and  phase  distortion  introduced  at  the  in- 
put of  the  preamplifier  by  the  network  used  to  couple  the  output  of  the  pick-up  tube  to  the 
grid  of  the  first  stage  of  the  preamplifier.  Use  of  a  high  value  of  load  resistor  (30,000  to 
100,000  ohms)  compared  to  the  value  which  would  be  chosen  for  flat  response  (1000  to 


Table  1.    Flexible  Cable  Used  between  Television  Cameras  and  Equipment  Racks 


Conductor 

Cable  Mfrs.  Maximum 

A.W.G. 

No. 

Function 

No. 

Color 

E 
(AC) 

E 
(DC) 

I 

(Amps) 

Orthicon 

Iconoscope 

Image  Orthicon 

I 

White 

250 

450 

2.4 

20 

Mult,  focus  +150 
volts  to  +250  volts 

Ike  focus  —350  volts 
to  -600  volts 

Mult,  focus  +T50 
volts  to  +250  volts 

2 
3 

Bkck 

Ring   —50  volts  to 
+150  volts 

Decelerator  —50  volts 
to  +150  volts 

Yellow 

1000 

1400 

Ike  grid  —950  volts 
to  —1150  volts 

Image  focus  200  volts 
to  —1000  volts 

4 
_ 

Red 

2000 

2800 

Multiplier  +1000 
volts  to  +2000  volts 

Ike  cathode  -1000 
volts  to  —1200  volts 

Multiplier  +1000 
volts  to  +2000  volts 

Black 

4.0 

18 

1  10  volts  a.c. 

110  volts  a.c. 

110  volts  a.c. 

6 
7 

White 

Red 

250 

450 

2.4 

20 

Bias  -105  volts 

Bias  —105  volts 

Bias  —105  volts 

8 

Brown 

Video  +51  +280  volts 

Video  +B  +280  volts 

Video  +5  +280  volts 

9 

Orange 

Deflection  +  B 

Deflection  +  B 

TO 

White 

50 

70 

Horizontal  center 

Edge  light 

Horiz.  center 

11 

Black 

Back  light 

12 

Red 

Target  —15  volts  to 
+15  volts 

Target  —15  volts  to 
+15  volts 

13 

Green. 

250 

450 

Beam  0  to  —45  volts 

Beam  0  to  —45  volts 

14 
_ 

Black 

Signal  light 

Signal  light 

,  Signal  light 

Vertical  deflection 

Vertical  deflection 

Vertical  deflection 

16 

Red 

17 

White 

Program  phone 

Program  phone 

Program  phona 

18 

Red 

19 

Blue 

Wall  focus  +100  volts 
to  +250  volts 

Wall  focus  +100  volts 
to  +250  volts 

20 

Green 

50 

70 

Alignment  field 

Alignment  field 

21 

Black 

250 

450 

Phone 

Phone 

Phone 

22 

Green 

23 
24 

White 

Front  focus  field 

Front  focus  field 

Yellow 

25 

Blue 

50 

70 

26 
27 

Brown 

600 

850 

Horizontal  S.T. 

Horn,  deflect. 

Horiz.  driving 

Orange 

Blanking 

Blanking 

Blanking 

28 

;  Yellow 

300 

500 

Video 

Video 

Video 

29 

Blue 

30 

Yellow 

50 

70 

Rear  focus  field 

Rear  focus  field 

31 

Brown 

32 

Orange 

Alignment  field 

Alignment  field 

20-24 


STUDIO  CAMERA  DESIGN 


20-25 


3000  ohms)  offers  advantages  in  discrimination  against  low-frequency  noise  and  micro- 
phonic  disturbances  in  the  first  preamplifier  stage,  because  it  results  in  a  relatively  greater 
level  of  low-frequency  signal  being  applied  to  the  first  grid  than  if  a  flat  system  is  used. 
On  the  other  hand,  tubes  having  a  self-contained  electron  signal  multiplier  such  as  the 
Image  Dissector  or  the  Image  Orthicon  can  provide  a  relatively  large  signal  output  current 
so  that  a  relatively  low  value  of  signal  load  resistor  can  deliver  sufficient  signal  voltage,  m 
many  cases,  to  override  noise  and  microphonic  disturbances  generated  in  the  first  therm- 
ionic amplifier  stage. 

"When  a  high  value  of  signal  load  resistor  is  used,  it  is  the  practice  to  compensate  for  the 
frequency  and  phase  distortion  introduced  by  incorporating  a  "high-peaker"  stage  in  the 
video  preamplifier.  Figure  2  is  a  block  diagram  illustrating  one  form  of  high-peaking. 


Pickup  tube 

(EZ» 

Load  resfstor- 


6-i-~      £  + 
FIG.  2.     High  Peaker  Stage  Employing  Resistance  and  Inductance  in  the  Plate  Circuit 

The  load  impedance  Z\  for  the  pick-up  tube  consists  of  Ri  and  Ci  in  parallel,  d  being  the 
combined  output  capacitance  of  the  pick-up  tube,  stray  and  wiring  capacitances,  and  the 
input  capacitance  of  the  first  video  amplifier  stage.  The  video  amplifier  stages  following 
the  input  circuit  are  designed  for  uniform  response  over  the  required  band  width.  These 
feed  a  high-peaker  stage  Vi  having  an  effective  plate  load  impedance  Z%  at  high  frequencies 
consisting  of  R%  and  Z/2  in  series  (neglecting  £2) .  Since  the  internal  impedance  of  both  the 
pick-up  tube  and  Vi  will  be  relatively  high  (they  may  be  considered  as  constant-current 
generators)  and  since  the  amplifier  between  the  pick-up  tube  and  V\  is  designed  for  uni- 
form response,  the  overall  response  of  the  system  is  proportional  to  the  product  ZiZ%.  If 
the  values  of  Z\  and  Zz  are  chosen  so  that  ZiZ%  =  A?  =  constant  (inverse  networks) ,  an 


Normal  Interstage 
7  shunt  peaking 


i 

100,000 

82 

i 
£  10-110 

: 

/ 

5 

|     Normal  low-f 
I—  ,&?  compens* 

=  2  MF 

6800     =; 

•=•                                       High-peaking 
adjustment 

(.280  v  O 

« 
• 
< 
< 
• 

f 

Fia.  3.     High  Peaker  Stage  Utilizing  Variable  Cathode  By-pass 

overall  flat  response  will  be  obtained  for  frequencies  below  the  point  where  the  resonance 
of  Ca  with  1/2  begins  to  have  important  effect.  This  condition  is  fulfilled  when  R^C\  = 
LZ/RZ.  It  has  been  found  that,  if  Z/2  is  chosen  such  that  the  resonance  of  Ca  with  L%  occurs 
at  a  frequency  about  1.5  or  more  times  higher  than  the  top  video  frequency  for  which  the 
balance  of  the  amplifier  is  designed,  the  effect  of  C2  can  be  neglected. 

Obviously,  by  adding  a  series  coil  to  the  input  circuit,  a  pair  of  inverse  networks  could 
be  provided  which  would  exactly  compensate  for  the  distortion  including  the  effect  of  Cj. 


20-26 


TELEVISION 


From  a  practical  standpoint,  however,  the  simpler  arrangement  of  Fig.  2  has  been  found 
to  give  satisfactory  results.  Two  other  high-peaker  circuits  which  are  simple  and  yet  can 
be  designed  to  give  satisfactory  compensation  over  a  reasonably  wide  band  are  illustrated 
in  Figs.  3  and  4.  A  variety  of  other  circuits  and  compensation  methods  can  also  be  used. 
The  low-frequency  compensation  of  the  preamplifier  generally  follows  standard  practice, 
the  amplifier  being  designed  to  pass  without  appreciable  distortion  a  60-cycle  square  wave 
applied  to  the  input  terminals.  In  some  designs  of  Iconoscope  preamplifiers,  an  additional 
network  consisting  of  150,000  to  300,000  ohms  shunted  by  0.001  to  0.005  pi  is  placed  in 
series  with  the  low-potential  end  of  the  signal  load  resistor  to  obtain  improved  operation 


6AC7 


3900 


+  280  v  O- 


FIG.  4.     High  Peakcr  with  Variable  Series  Capacitor 


at  low  and  medium  frequencies.  When  this  is  done,  an  appropriate  network  must  be  in- 
serted in  one  of  the  later  stages  of  the  amplifying  system  to  compensate  for  this  low- 
frequency  pre-emphasis. 

13.  STUDIO  EQUIPMENT 

CONTROL  ROOM.  In  a  typical  studio  control  room  for  Iconoscope  cameras,  the 
rack  equipment  for  each  camera  chain  consists  of  the  following: 

(a)  Horizontal  deflection  sawtooth  wave  generator,  with  the  output  modulated  by 
sawtooth  waves  of  vertical  frequency  for  keystoning.  The  amplitude  of  the  sawtooth 
wave,  as  well  as  the  amount  of  keystoning,  is  adjustable.  An  output  transformer  feeds 
the  low-impedance  deflection  coils  in  the  camera  via  a  75-ohm  coaxial  cable.  An  im- 
pedance step-up  transformer  located  in  the  camera  is  sometimes  used  to  feed  the  de- 
flection coils. 

(Z>)  Vertical  deflection  sawtooth  wave  generator  with  an  output  transformer  feeding  a 
75-ohm  coaxial  line  which  connects  to  the  camera  vertical  deflection  coils.  An  impedance 
step-up  transformer  located  in  the  camera  is  sometimes  used  to  feed  the  deflection  coils. 

(c)  Regulated  power  supplies  for  deflection  amplifiers,  video  amplifiers,  and  the  Icono- 
scope. 

(d)  Studio  amplifier.     This  amplifier  provides  video  Outputs  for  feeding  monitoring 
circuits  and  the  master  control  room.     The  output  circuits  feed  75-ohrn  coaxial  lines  at  a 
signal  voltage  level  between  0.5  volt  and  1.0  volt  peak  to  peak,  depending  on  the  standard 
level  adopted  for  a  given  plant.     The  output  of  the  studio  amplifier  contains  picture 
blanking  signals  but  not  synchronizing  signals.     An  excess  of  picture  blanking  signals  is 
introduced  in  one  of  the  later  stages  of  the  studio  amplifier  followed  by  an  adjustable 
clipper  or  pedestal  (brightness)  control.     The  amplifier  is  provided  with  a  video  gain  con- 
trol. 

(e)  Shading  amplifier.     Certain  types  of  camera  tubes,  including  the  Iconoscope,  gen- 
erate a  spurious  signal  output  in  addition  to  the  signals  representative  of  picture  informa- 
tion.    The  effect  on  the  television  picture  of  uncompensated  spurious  signals  is  to  produce 
shaded  areas  not  present  in  the  original  scene. 


STUDIO  EQUIPMENT  20-27 

It  has  been  found  in  practice  that  the  effect  of  spurious  signals  can  readily  be  compen- 
sated by  the  addition  of  a  few  simple  wave  forms  of  appropriate  amplitude  and  polarity  to 
the  camera  video  output.  Since  the  shading  wave  forms  and  amplitudes  required  vary 
with  scene  illumination,  it  is  the  practice  to  provide  individual  manual  adjustment  of 
shading  waveforms  for  each  Iconoscope  camera.  Pick-up  tubes  which  do  not  always  re- 
quire shading  signals  are  the  Image  Dissector,  the  RCA  type  1840  Orthicon,  and  the 
Image  Orthicen. 

The  following  shading  wave  forms,  adjustable  in  amplitude  and  polarity,  are  provided 
for  each  camera  chain:  (1)  horizontal  sawtooth;  (2)  vertical  sawtooth;  (3)  horizontal 
parabola  (adjustable  clipping);  (4)  vertical  parabola  (adjustable  clipping);  (5)  vertical 
sine  wave  with  adjustable  phase  (the  inclusion  of  vertical  sine  wave  is  optional). 

When  a  studio  is  equipped  with  more  than  one  Iconoscope  camera,  it  is  usually  econom- 
ical to  provide  a  shading  generator  having  a  low-impedance  pushpull  output  bus  for  each 
shading  wave  form.  Relatively  high-impedance  center-tapped  potentiometers  can  then 
be  connected  in  parallel  with  these  buses  to  feed  a  shading  isolation  and  mixing  amplifier, 
associated  with  each  camera  chain. 

Shading  signals  should  be  added  to  the  video  signal  in  a  low-level  stage  of  the  video  sys- 
tem prior  to  the  introduction  of  picture  blanking.  Introduction  of  shading  into  a  low- 
level  stage  (it  is  sometimes  fed  into  an  early  stage  of  the  preamplifier  in  the  camera)  tends 
to  prevent  overload  of  the  higher-level  stages  due  to  uncompenaated  tilt  in  the  video  signal 
wave  form. 

The  following  operating  controls  should  be  located  on  the  studio  console  for  each  camera 
chain:  (1)  Iconoscope  beam  current;  (2)  Iconoscope  focus;  (3)  video  gain  (contrast);  (4) 
pedestal  (brightness) ;  (5)  all  shading  controls. 

The  following  controls  which  generally  require  attention  only  during  initial  warm-up 
of  the  equipment  need  not  be  mounted  on  the  console  but  should  be  within  easy  reach  of 
the  video  operator:  (1)  vertical  deflection  amplitude;  (2)  vertical  deflection  centering; 
(3)  keystone  adjustment;  (4)  horizontal  deflection  amplitude;  (5)  horizontal  deflection 
centering;  (6)  bias  light  adjustment. 

Though  the  example  given  above  deals  with  Iconoscope  cameras,  the  same  general 
principles  and  layout  apply  for  other  types  of  pick-up  tubes,  except,  of  course,  that  dif- 
ferent arrangements  of  operating  controls  and  adjustments  are  necessary.  The  need  for 
shading  signals  is  generally  associated  with  Iconoscopes  alone,  but  availability  of  at  least 
sawtooth  and  parabola  wave  forms  is  often  advantageous  when  using  other  types  of  camera 
tubes  which  nominally  do  not  require  it.  Thus  a  tube  with  somewhat  non-uniform  sen- 
sitivity over  different  parts  of  its  photoelement  may  produce  a  satisfactory  picture  if 
shading  signals  are  available. 

AMPLIFIER  DESIGN.  In  the  usual  equipment  layout,  the  studio  amplifier  performs 
the  following  functions: 

1.  Provides  for  the  insertion  of  shading  wave  forms  if  they  have  not  previously  been 
added  in  the  preamplifier. 

2.  Provides  for  the  addition  of  picture  blanking  signals  of  adjustable  amplitude  to  the 
video  signal. 

3.  Provides  for  variable  gain  control  (automatic  gain  control  is  sometimes  used)  of  the 
video  signal. 

4.  May  include  one  or  more  of  the  following  corrective  features:  (a)  gamma  correction; 
(b)  aperture  distortion  correction;  (c)  correction  for  phase  distortion. 

5.  The  studio  amplifier  usually  provides  at  least  two  independent  outputs  of  combined 
video  and  blanking  signals  at  levels  of  0.5  volt  to  1.0  volt  peak  to  peak  at  75-ohm  im- 
pedance.    One  of  these  outputs  normally  feeds  the  master  control  and  switching  point 
where  the  studio  output  may  be  switched  to  the  transmitter.     The  other  output  feeds  the 
local  studio  picture  monitoring  circuits. 

The  video  coupling  networks  used  in  the  studio  amplifier  follow  the  conventional  de- 
signs. Shading  signals  are  generally  added  via  a  tube  whose  plate  or  cathode  feeds  an 
impedance  common  to  a  plate  or  cathode  of  one  of  the  video  stages.  Blanking  signals 
are  generally  added  through  a  tube  whose  plate  feeds  an  impedance  common  to  the  plate 
of  one  of  the  later  video  amplifying  stages.  An  excess  of  picture  blanking  signal  is  added, 
being  clipped  off  by  a  variable  clipping  arrangement  in  the  following  video  stage.  Care 
should  be  exercised  in  the  design  to  provide  an  exceptionally  linear  clipping  arrangement 
to  avoid  crowding  of  signal  voltages  corresponding  to  the  darker  portions  of  the  picture. 

MONITORING  AND  SWITCHING  FACILITIES.  Studio  monitoring  facilities  gen- 
erally consist  of  a  picture  monitor  showing  the  outgoing  picture  plus  one  picture  preview 
monitor  for  each  camera  in  operation  in  the  studio.  The  outgoing  picture  or  "on  the  air" 
monitor  should  preferably  be  fed  through  a  return  feed  from  the  master  control  point  and 
should  be  synchronized  from  the  synchronizing  signals  which  have  been  added  to  the  video 


20-28 


TELEVISION 


and  blanking  components  at  the  master  control  point.  The  preview  monitors  are  gen- 
erally of  the  "driven  type,"  i.e.,  locked-in  by  the  horizontal  and  vertical  driving  pulses 
utilized  to  operate  the  camera  deflection  circuits.  When  this  is  done,  the  preview  monitors 
may  be  switched  from  one  camera  output  to  another  without  loss  of  synchronization. 
Also,  since  the  camera  outputs  normally  do  not  provide  synchronizing  signals,  the  use  of 
driving  impulses  gives  more  stable  results  than  blanking  signals  for  preview  monitor 
synchronization.  The  on-the-air  monitor  should  be  synchronized  from  pulses  contained 
in  the  composite  input  signal,  however,  because  it  is  often  necessary  to  connect  it  to  view 
pictures  supplied  from  other  sources  in  the  normal  course  of  operations. 

One  satisfactory  method  of  switching  the  video  monitors  is  to  provide  them  with  high- 
impedance  inputs  (via  cathode  followers  in  some  designs)  so  that  one  or  more  can  be 
switched  to  the  75-ohm  monitoring  output  bus  of  any  one  of  the  studio  amplifiers.  Mon- 
itor switching  may  be  accomplished  by  mechanical  switches,  by  switching  relays,  or  by 
electronic  means.  A  relay  switching  system  controlled  by  a  set  of  pushbuttons  and  in- 
dicator lamps  mounted  on  the  control  console  is  sometimes  used. 

In  addition  to  picture  monitoring  facilities,  wave-form  monitors  have  to  be  provided 
as  a  continuous  check  on  the  shading  and  voltage  level  of  the  signal  generated  by  each  cam- 
era chain  as  well  as  that  being  fed  out  of  the  studio.  A  pushbutton-controlled  switching 
system  for  the  wave-form  monitors  is  desirable. 

TIME  DELAY  NETWORKS.  In  the  usual  television  studio  plant  having  more  than 
one  studio  or  signal  source,  it  is  the  practice  to  add  the  synchronizing  signals  to  the  video 


R— "sir 


0<1^8R  =s  self-lad ucta nee  of  each  coll. 


M=  -—:  —  s  mutual  Inductance  of  LI  and  La.   (  Colls  are  wound  so-  that  the  fields  are  series  aiding,  thus  gfvFnff  an 
effective  negative  mutual  Inductance). 


/—top  frequency  required  to  reproduce  original  pulse. 

R  =  deslgn  Impedance. 

Approximate  delay  per  section  —"^  —  seconds. 

The  LI,  L2.  M  assembly  Is  a  single-layer  coil  with  a  center  tap,  the  wire  size  spacing  between  turns  and  diameter 
of  form  being  so  chosen  that  the  following  conditions  are  satisfied. 


A  center  tapped  coll  satisfying  these  conditions  Is  the 
equivalent  of  two  colls  each  of  seff-Inductance  Li  and 
of  negative  mutual  inductance  M 


,            U-M 
— 


FIG.  5.     Delay  Network  Employing  Negative  Mutual  Inductance 

and  blanking  signals  at  one  point  in  the  system,  usually  the  master  switching  or  control 
point.  This  is  done  for  the  following  reasons:  (1)  failure  of  the  studio  output  does  not 
disturb  the  transmitter  or  throw  receivers  out  of  synchronization;  (2)  synchronizing  signal 
level  fed  to  the  transmitter  can  be  made  independent  of  variations  in  studio  output  level; 
(3)  insertion  of  synchronizing  signals  in  the  system  at  a  single  point  just  prior  to  the  studio- 
transmitter  link  is  the  most  practical  way  to  maintain  synchronizing  signals  at  the  exact 
level  required  by  the  transmitter. 

Since  the  cables  connecting  the  various  studios  with  the  master  control  position  are 
usually  of  different  lengths,  electrical  networks  are  employed  to  delay  the  signals  arriving 
via  shorter  cables  so  that  the  proper  "front  porch"  margin  between  the  starting  time  of  the 
horizontal  picture  blanking  pulses  and  the  starting  time  of  the  horizontal  synchronizing 
pulses  will  be  maintained  when  synchronizing  signals  are  added.  The  delay  networks 
usually  consist  of  a  number  of  sections  of  artificial  line  utilizing  lumped  constants.  Though 
it  would  be  possible  to  insert  the  delay  networks  in  the  studio  video  outputs,  it  is  gen- 


GAMMA  (TRANSFER  CHARACTERISTIC)  20-29 

erally  preferable  to  insert  them  in  the  coaxial  lines  feeding  the  studio  with  the  camera  driv- 
ing, camera  blanking,  and  picture  blanking  signals.  This  is  more  economical  as  less  perfect 
delay  networks  can  be  used  and  the  effects  of  any  slight  transmission  line  reflections  due 
to  incorrect  termination  can  readily  be  removed  by  clipper  stages  in  the  studio  deflection 
and  blanking  amplifiers. 

DESIGN  OF  DELAY  NETWORKS.  Studio  delay  networks  may  be  designed  in  a 
number  of  forms.  ^  A  common  type  is  an  artificial  line  with  Z0  =  75  ohms  in  the  form  of  a 
ladder  network  with  a  number  of  series  coils  and  shunt  capacitors.  This  type  of  artificial 
line  may  be  treated  as  a  low-pass  filter  with  the  cut-off  chosen  somewhat  higher  than  the 
top  operating  frequency.  As  in  usual  filter  practice,  this  type  of  network  is  best  ter- 
minated by  a  double  M-derived  network  at  its  output.  A  design  using  negative  mutual 
inductance  which  gives  excellent  performance  is  shown  in  Fig.  5.  The  top  operating  fre- 
quency may  be  chosen  on  the  basis  of  the  number  of  harmonics  of  line-scanning  frequency 
required  to  reproduce  a  pulse  of  a  given  steepness.  The  order  n  of  the  highest  harmonic 
required  is  n  =  lOO/p,  where  p  is  the  time  of  rise  of  the  wave  expressed  in  percentage  of 
the  fundamental  period  (reference  35) . 

Continuous  lengths  of  transmission  line  can  be  used  in  lieu  of  lumped  delay  networks. 
If  a  continuous  length  of  line  is  used  for  time  delay,  it  should  have  reasonably  low  attenua- 
tion and  uniform  time  delay  over  the  required  frequency  band. 

14.  GAMMA  (TRANSFER  CHARACTERISTIC) 

The  term  gamma  (7),  has  been  used  to  define  the  slope  of  the  curve  of  the  logarithm  of 
image  brightness  vs.  the  logarithm  of  object  brightness  (reference  36).  Unless  otherwise 
specified,  the  slope  7  refers  to  the  central  linear  portion  of  the  curve,  between  the  ex- 
tremities of  the  brightness  values  considered. 

Though  the  term  gamma  is  a  useful  concept  when  making  comparisons  with  photog- 
raphy, the  trend  is  away  from  it  in  connection  with  television  systems.  As  a  matter  of 
fact,  the  term  must  be  applied  with  caution  since  it  often  leads  to  erroneous  conclusions. 
The  reason  for  this  is  twofold.  First,  gamma,  being  a  numeric  and  referring  only  to  the 
central  portion  of  the  characteristic,  tells  nothing  about  the  effect  of  the  shape  of  the  toe 
and  knee  of  the  curve;  second,  the  concept  is  difficult  to  apply  in  many  portions  of  the 
television  system  where  a  reference  to  picture  black  is  not  directly  available.  Therefore, 
the  term  gamma,  if  used  at  all,  should  be  restricted  to  comparisons  of  original  scene  bright- 
nesses and  final  image  brightnesses  in  an  overall  sense. 

"Transfer  characteristic' '„  (reference  28)  is  the  name  that  has  been  given  to  the  logarith- 
mic plot  of  light  input  vs.  signal  voltage  output  of  a  television  transducer.  Thus,  the 
Iconoscope  performance  curve  of  Fig.  1,  plotted  to  logarithmic  coordinates,  could  be  called 
the  "signal  current  vs.  illumination  transfer  characteristic"  of  an  Iconoscope. 

GAMMA  CORRECTION.  It  can  be  shown  that  the  overall  signal-to-noise  ratio  of  a 
television  pick-up,  transmitting,  and  receiving  system  is  improved  by  transmitting  a 
logarithmic  light  vs.  voltage  characteristic.  Though  the  exact  shape  of  the  curve  has  not 
been,  standardized  (1949),  it  appears  to  be  accepted  that  the  light  vs  voltage  character- 
istic of  the  Iconoscope  (see  Fag.  1)  is  close  to  the  ideal  shape.  The  reproducer  normally 
has  the  opposite  curvature  to  Fig.  1.  This  is  conveniently  obtained  by  providing  a  picture 
tube  with  the  required  light  output  vs.  grid  voltage  transfer  characteristic. 

When  a  pick-up  tube  such  as  an  Image  Dissector  or  an  RCA  type  1840  Orthicon,  having 
linear  voltage  output  vs.  illumination  transfer  characteristic,  is  used,  it  is  necessary  to 
incorporate  a  non-linear  element  in  the  amplifying  system  to  achieve  the  desired  logarith- 
mic curvature.  An  amplifier  designed  for  this  purpose  is  called  a  gam  ma-correction 
amplifier. 

A  wide  variety  of  circuits  may  be  used  for  gamma  correction.  The  following  design 
principles  apply:  (a)  Black  level  must  be  established  at  the  input  to  correction  stage  so 
that  the  correction  can  be  applied  independent  of  the  excursions  of  the  AC  axis  of  the  video 
signal  with  respect  to  black  level.  (6)  The  correction  should  be  applied  prior  to  the  video 
stage  in  which  picture  blanking  pedestals  are  inserted,  (c)  The  correction  amplifier  should 
be  designed  so  that  the  amount  of  correction  applied  is  variable  between  no  correction  at 
all  and  the  maximum  value.  The  maximum  value  is  limited  by  the  minimum  signal-to- 
noise  ratio  that  can  be  tolerated.  A  difference  in  video  gain  between  the  signals  corre- 
sponding to  the  dark  portions  and  the  light  portions  of  the  picture  of  10  :  1  or  even  100  :  1 
is  often  desirable.  A  gain  difference  of  10  times  is  the  most  that  can  be  used  in  many 
cases,  however,  owing  to  signal-to-noise  ratio  limitations.  The  amount  of  correction  to 
be  applied  for  a  given  soene  is  subject  to  artistic  as  well  as  technical  requirements,  (d) 
For  a  television,  system  capable  of  reproducing  only  a  relatively  narrow  contrast  range,  the- 


20-30 


TELEVISION 


exact  shape  of  the  gamma-correction  curve  is  relatively  unimportant  as  far  as  the  eye  is 
concerned  when  viewing  the  overall  result. 

A  common  type  of  gamma-correction  amplifier  utilizing  tubes  of  different  character- 
istics in  parallel  is  shown  in  Fig.  6.  Another  system  having  a  diode  connected  across  the 
cathode  resistor  of  a  video  stage  is  shown  in  Fig.  7. 


.6  v  to  10  v 

peak  to  peak 
6AC7  50 


Video  output 

IRj 

>— Black  level 

_rfffftf\_|| — 


~~ 

7  adjustment  changing 
screen  voltage  from 
+25  v  to  +150  v 
FIG.  6.     Gamma  Correction  Stage  Using  Tubes  in  Parallel 


White, 


— 105  v 

Note:  Ri  controls  change  In  ' 
R2  controls  signal  level 
at  which  change  In  7 
occurs 
R3  controls  black-level  clipping 

at  Input 
FIG.  7.     Gamma  Correction  Stage  with  Variable  Cathode  Degeneration 


15.  APERTURE  CORRECTION 

When  the  size  of  the  pick-up  or  reproducer  scanning  spot  is  appreciable  compared  to  a 
picture  element,  aperture  distortion  takes  place.  For  a  mechanical  scanning  system  or 
an  equivalent  system  in  which  the  dimensions  of  the  scanning  spot  are  accurately  known, 
the  amplitude  vs.  frequency  distortion  caused  by  the  finite  size  of  the  aperture  may  be 


FILM  PICK-UP  20-31 

accurately  calculated  (reference  6).  For  cathode-ray  systems  the  effects  of  aperture  dis- 
tortion are  best  determined  by  test.  In  a  pick-up  tube  using  cathode-ray  beam  scanning, 
for  example,  the  effects  of  aperture  distortion  may  be  measured  by  determining  the  signal 
output  obtained  by  scanning  different  patterns  of  uniformly  illuminated  alternate  black 
and  white  lines  of  various  pitches.  A  measured  signal  output  vs.  picture  element  size 
curve  having  been  obtained  for  a  given  type  pick-up  tube,  an  amplitude  correction  equal- 
izing network  may  be  designed  to  compensate  for  the  aperture  distortion  (reference  2). 
The  complete  aperture  correction  equalizer  should  include  a  phase  equalizer  to  compensate 
for  any  phase  distortion  introduced  by  the  amplitude  correction  network.  Apertures  of 
irregular  shape  may  also  introduce  phase  distortion.  The  aperture  correction  network, 
if  designed  for  a  characteristic  impedance  of  75  ohms,  may  be  connected  between  the  studio 
amplifier  output  and  the  coaxial  line  feeding  the  master  control  position. 

The  amount  of  aperture  correction  that  can  be  applied  in  a  given  case  is  limited  by  the 
extent  to  which  a  corresponding  reduction  in  the  signal-to-noise  ratio  of  the  studio  output 
can  be  tolerated.  As  a  practical  matter,  it  is  found  that  the  inherent  resolution  capa- 
bilities of  most  types  of  television  pick-up  tubes  is  great  enough  so  that  a  special  network 
for  aperture  correction  is  unnecessary.  Adjustment  of  the  high-peaker  stage  (see  article 
12)  for  the  sharpest  picture  may  often  include  some  inadvertent  correction  for  aperture- 
distortion  effects. 

16.  FILM  PICK-UP 

To  assure  program  continuity  and  flexibility  the  following  may  be  considered  the  min- 
imum equipment  requirements  for  the  television  film  pick-up  studio:  (1)  two  television 
film  pick-up  cameras  with  associated  control,  monitoring,  and  amplifying  equipment;  (2) 
two  35-mm  sound-mo tion-picture  projectors;  (3)  two  16-mm  sound-motion-picture  pro- 
jectors; (4)  two  35-mm  slide  projectors;  (5)  sound  control  and  monitoring  facilities. 

The  amplifying,  control,  and  monitoring  equipment  for  the  film  pick-up  studio  is  es- 
sentially the  same  as  that  for  the  direct  pick-up  studio.  Additional  console  controls  re- 
quired are  remote  projector  starting  and  phasing.  Since  the  synchronizing  generator  is 
generally  locked-in  with  the  local  power  mains,  synchronous  motors  are  used  to  operate  the 
film  projectors.  A  convenient  method  of  phasing  the  projectors  is  to  shift  the  phase  of  the 
synchronizing-generator  60-cycle  reference  voltage,  thereby  shifting  the  phase  of  camera 
scanning  with  respect  to  the  projector.  Mechanical  phasing  methods  are  also  satisfactory. 

Both  storage  and  non-storage  types  of  pick-up  tubes  are  suitable  for  television  film 
pick-up.  Since  the  standard  sound  film  frame  rate  is  24  per  second  and  the  standard  tele- 
vision frame  rate  is  30  per  second,  the  difference  is  made  up  by  scanning  consecutive 
frames  2  and  3  times  respectively  at  the  60-cycle  interlaced  field  deflection  rate. 

In  the  case  of  non-storage-type  pick-up  tubes,  such  as  the  Image  Dissector,  the  optical 
image  of  the  film  or  an  appropriate  section  of  it  must  be  projected  optically  on  the  photo- 
cathode  at  the  time  scanning  takes  place.  Intermittent  film  projection  is  not  considered 
practical  for  non-storage  tubes  since  the  film  pull-down  time  is  limited  to  approximately 
0.00117  second  according  to  present  standards.  Special  projection  methods  are  therefore 
required.  Some  of  these  employ  uniform  motion  of  the  film  in  conjunction  with  optical 
and  electronic  means  to  provide  scanning  of  alternate  film,  frames  in  the  required  2,3 
sequence  (reference  37). 

For  storage-type  pick-up  tubes,  such  as  the  Iconoscope,  the  optical  image  is  flashed  on 
the  mosaic  during  vertical  blanking  time  and  the  stored  "charge-image"  scanned  off  dur- 
ing the  normal  vertical  scanning  time  (reference  38) .  The  film  is  moved  during  the  ver- 
tical scanning  time  when  the  mosaic  is  dark.  For  35-mm  film  a  special  intermittent  mo- 
tion designed  so  that  the  interval  between  pull-downs  alternates  between  1/20  and  1/30 
second  may  be  used  to  attain  the  required  projection  sequence,  the  maximum  film  pull- 
down time  available  being  approximately  0.015  second  under  1946  standards.  Projectors 
for  16-mm  film  can  be  designed  for  a  pull-down  time  of  0.007  second  or  less,  in  which  case 
a  normal  equally  spaced  intermittent  may  be  used.  A  synchronously  driven  rotating 
shutter  placed  between  the  projector  lens  and  the  pick-up  tube  is  used  to  flash  the  picture 
on  the  mosaic.  Proper  phasing  between  the  light  flashes  and  the  scanning  may  be  obtained 
by  mechanical  or  electrical  means. 

Automatic  brightness  control  (automatic  adjustment  of  the  picture  blanking  clipper 
bias  in  the  studio  amplifier)  may  be  provided  for  film  pick-up  by  means  of  an  auxiliary 
photocell  to  pick  up  the  light  from  the  film  a  few  frames  ahead  of  projection.  This  photo- 
cell output  is  integrated  and  applied  to  the  blanking  clipper  bias  circuit. 


20-32 


TELEVISION 


17.  MASTER  CONTROL  POSITION 

In  the  usual  television  plant,  where  there  are  several  signal  sources,  including  remote 
pick-ups,  all  switching  and  the  final  monitoring  of  the  signal  before  it  is  sent  to  the  trans- 
mitter are  done  at  a  central  point  called  the  master  control  position.  The  synchronizing 
generator,  the  synchronizing  signal  distribution  amplifiers,  the  video  line  amplifiers,  and 
the  various  items  of  test  equipment  used  in  connection  with  operations  are  usually  located 
in  a  main  equipment  room  adjacent  to  the  master  control  position. 

A  typical  studio  master  control  pulse  and  video  system  layout  is  shown  in  Fig.  8.  The 
synchronizing  generator  outputs  are  fed  to  the  inputs  of  a  number  of  pulse  distribution 


I        DIRECT  PICK-UP  STUDIO 

CAMERA 


DIRECT   STUDIO   CAMERA  RELAYS 
PUSHBUTTON    CONTROLS    IN 
STUDIO  CONTROL  CON-SOLE 


Main  equipment  room 

VIDEO   LINE 


FIG.  8.     Block  Diagram  Showing  Pulse  Distribution  to  Studios  and  Video  Switching  Circuits  at  the 

Master  Control  Position 

amplifiers  as  shown.  A  separate  pulse  distribution  amplifier  feeds  each  studio  and  the 
monoscope.  A  separate  distribution  amplifier  is  also  reserved  for  synchronizing  signals. 
This  system  of  pulse  distribution  isolates  one  studio  from  another  so  that  an  accidental 
short  circuit  or  failure  in  a  pulse  line  to  one  studio  cannot  affect  the  others.  A  spare  pulse 
distribution  amplifier  is  held  in  readiness  in  case  of  failure  of  one  of  the  amplifiers.  It  is 
also  good  practice  to  have  a  spare  synchronizing  generator  in  operation  at  all  times  and  to 
provide  means  for  quickly  switching  it  into  service  if  the  regular  generator  fails. 

As  shown  in  the  diagram,  delay  networks  are  inserted  in  the  pulse  lines  to  all  studios 
except  the  one  having  the  longest  cable  run.  The  delay  networks  are  adjusted  (by  adding 
or  removing  sections)  so  that  the  time  of  arrival  of  the  blanking  signals  from  each  studio 
at  the  master  control  position  switching  relays  will  be  the  same  and  will  be  such  that  the 
proper  front  porch  delay  of  synchronizing  signals  will  be  maintained  when  the  signals  are 
added.  There  is  generally  enough  coincidental  time  delay  in  the  camera  vertical  deflection 
system  so  that  no  delay  networks  are  required  to  time  the  vertical  pulses  properly.  It  will 
be  noted  that  the  output  of  each  studio  camera  is  brought  to  a  bank  of  switching  relays  as- 
sociated with  that  studio  and  located  at  the  master  control  position.  These  relay  switch- 
ing banks  in  turn  feed  isolation  amplifiers  which  are  arranged  to  feed  banks  of  relays  con- 
necting the  studio  outputs  to  the  line  amplifier  inputs.  Each  line  amplifier  provides  two 


PULSE  MEASUREMENTS  20-33 

outputs,  one  of  which  can  be  patched  to  feed  the  transmitter  and  the  other  can  provide  a 
monitoring  return-feed  to  one  of  the  studios. 

Normally,  only  one  or  two  of  the  amplifiers  (regular  and  spare)  will  be  used  for  feeding 
the  picture  transmitter.  The  other  line  amplifiers  are  patched  to  provide  monitoring  feeds 
for  the  studio  control  rooms,  announcers'  booths,  clients'  booths,  or  other  points  in  the 
television  plant  where  line-fed  monitors  are  used.  As  indicated,  coaxial  patch  cords  in 
connection  with  coaxial  jacks  are  provided  so  that  changes  in  system  interconnections  can 
be  rapidly  effected  as  dictated  by  operating  requirements.  In  addition,  a  number  of  spare 
coaxial  cables  connecting  to  each  studio  terminate  in  a  jack-field  adjacent  to  the  master 
control  console.  This  makes  it  possible  to  patch  up  special  circuit  arrangements  as  re- 
quired. Means  are  also  provided  for  patching  up  special  control  circuits.  For  example, 
the  effects  studio  output,  which  may  be  a  picture  of  a  title  slide,  can  be  patched  into  one 
of  the  spare  camera  relays  in  the  relay  bank  associated  with  the  direct  pick-up  studio. 
Thus  the  video  operator  in  the  direct  pick-up  studio  merely  presses  a  camera  switching 
button  when  he  wishes  to  insert  the  slide  in  the  outgoing  program. 

The  master  control  position  is  equipped  with  picture  and  wave-form  monitoring  facil- 
ities, a  television  receiver  used  as  an  off-the-air  monitor,  and  audio  monitoring  equipment. 
The  master  control  desk  is  normally  equipped  with  a  pushbutton-controlled  monitoring 
system  so  that  the  incoming  picture  from  any  of  the  sources  can  be  checked  as  required. 
A  wave-form  monitor  associated  with  the  picture  monitor  is  used  for  checking  signal  levels, 
pulse  wave  forms,  etc.  All  program  switching  between  the  various  signal  sources  and  the 
transmitter  is  done  at  the  master  control  position. 

A  control  unit  for  handling  remote  pick-ups  consisting  of  a  picture  monitor,  a  wave- 
form monitor,  and  a  variable  gain  video  amplifier  is  usually  provided.  The  output  of  this 
unit  feeds  into  one  of  the  relays  in  the  line  amplifier  bank  as  shown.  A  relay  is  also  pro- 
vided to  open  the  line  amplifier  synchronizing  signal  input  when  the  remote  relay  closes, 
as  the  remote  signal  normally  arrives  complete  with  synchronizing  signals.  When  the 
remote  synchronizing  generator  is  locked-in  with  the  same  power  supply  as  the  main 
generator,  it  is  possible  to  adjust  the  phase  of  the  two  synchronizing  signals  so  that  the 
rapid  switch  from  local  to  remote  pick-up  causes  no  appreciable  disturbance  of  receiver 
scanning. 

Conventional  relay  interlock  systems  are  used  to  prevent  closing  of  more  than  one  relay 
in  a  particular  bank  at  any  time  and  to  drop  out  one  relay  when  the  next  is  picked  up. 
Magnetic  delay  circuits  can  be  arranged  to  delay  the  drop-out  of  one  relay  until  just  before 
the  next  closes. 

A  wide  variety  of  electronic  fading  and  switching  systems  may  be  used  in  addition  to  the 
relay  system  shown.  Fades,  lap  dissolves,  superposition  of  one  image  on  another,  in 
whole  or  in  part,  may  be  accomplished  electronically. 

SYNCHRONIZING  GENERATOR.  Electronic  synchronizing  signal  generators, 
rather  than  electromechanical  generators,  are  almost  universally  used  for  providing  the 
pulses  needed  to  operate  the  television  plant.  The  type  generally  used  (reference  27) 
incorporates  an  AFC  oscillator  operating  at  double  line  frequency  in  connection  with  a 
frequency  divider  having  an  output  at  field  frequency.  The  output  of  a  discriminator 
comparing  the  local  60-cycle  power  frequency  with  the  output  of  the  frequency  divider 
controls  the  AFC  oscillator.  By  means  of  a  number  of  multivibrators,  delay  networks, 
and  pulse  shaping  networks,  the  generator  provides  the  following  outputs:  (1)  synchroniz- 
ing signals;  (2)  line-frequency  driving  pulses;  (3)  field-frequency  driving  pulses;  (4)  camera 
blanking  signals;  (5)  picture  blanking  signals.  Controls  are  provided  for  regulating  the 
duration  times  or  "widths"  of  the  various  pulses.  The  relative  starting  times  of  the 
various  pulses  are  usually  controlled  by  means  of  an  electrical  delay  network  incorporated 
within  the  unit.  The  outputs  of  the  synchronizing  generator  are  normally  available  at 
peak-to-peak  voltages  of  4  to  6  volts  at  an  impedance  of  75  ohms. 

The  general  practice  has  been  to  use  the  local  60-cycle  power  source  for  locking-in  the 
synchronizing  generator.  However,  some  designs  incorporate  a  selector  switch  for  locking 
the  synchronizing  generator  in  either  with  the  local  power  frequency  or  with  a  subharmonic 
of  a  crystal  oscillator.  Operation  of  the  synchronizing  generator  from  a  stabilized  fre- 
quency source  independent  of  the  local  60-cycle  power  mains  has  many  advantages. 

18.  PULSE  MEASUREMENTS 

In  order  to  insure  that  the  transmitted  blanking  and  synchronizing  pulse  wave  forms  con- 
form to  the  FCC  standards',  the  television  broadcasting  plant  must  be  provided  with  test 
equipment  for  measurement  of  the  relative  starting  times,  slopes,  and  duration  times  of 
these  pulses. 


20-34 


TELEVISION 


Space  does  not  permit  a  discussion  of  the  great  variety  of  measurement  techniques  that 
can  be  or  have  been  successfully  used.  However,  the  following  have  been  found  to  be 
convenient  and  are  sufficiently  accurate  for  practical  purposes  (reference  35) : 

SINE-WAVE  SWEEP.  A  method  that  allows  rapid  and  accurate  checks  to  be  made 
of  pulse  slopes,  widths,  and  delay  times  utilizes  an  oscilloscope  with  sine-wave  sweep  of 


110- 


140- 


FIG.  9.     Alignment  Chart  for  Pulse  Width,  Scope,  and  Relative  Time  Delay  Measurements  Using 

Sine  Wave  Horizontal  Sweep 

either  line  or  field  scanning  frequency  depending  on  the  character  of  the  pulse  to  be  meas- 
ured (reference  35).  The  line-frequency  sine  wave  is  obtained  by  filtering  out  the  fun- 
damental of  the  horizontal  pulses  from  either  the  synchronizing  signal,  the  picture  or 
camera  blanking  signals,  or  the  horizontal  driving  pulses.  For  accurate  results,  the  sine 
wave  used  for  measurement  purposes  should  be  well  filtered  so  that  the  arithmetic  sum  of 


OVERALL  VIDEO  SYSTEM  RESPONSE  20-35 

all ^  harmonics  does  not  exceed  1  per  cent.  The  local  60-cyele  a-c  mains,  filtered  as  re- 
quired, may  be  used  as  a  source  of  field-frequency  sine  wave  if  the  field  frequency  is  locked 
in  with  the  local  power  source.  The  oscilloscope  used  should  have  a  linear  horizontal 
sweep  width  of  about  100  mm  minimum.  Pulse  widths  are  measured  by  shifting  the  pulse 
to  the  center  of  the  screen  by  means  of  a  sweep  phase  shifter  and  measuring  the  dimensions 
indicated  on  Fig.  9  with  a  transparent  millimeter  scale  at  the  10  and  90  per  cent  amplitude 
points  of  the  wave. 

The  width  of  the  pulse  in  percentage  of  a  sine  wave  period  may  then  be  obtained  from 
the  nomographic  chart.  The  expanded  scale  should  be  used  for  widths  of  less  than  3  per 
cent.  Pulse  slopes  are  measured  by  shifting  the  pulse  edge  in  question  to  the  center  of  the 
screen  so  that  the  10  and  90  per  cent  amplitude  points  are  symmetrically  disposed  about 
the  center  line  of  the  sweep  and  scaling  C  or  CA  and  D.  The  C  dimension  appears  mag- 
nified compared  to  linear  sweep  when  sine  wave  is  used,  and  the  accuracy  of  measurement 
is  thereby  enhanced.  Pulse  delay  times  may  be  measured  with  respect  to  a  specific  time 
(such  as  the  starting  time  of  horizontal  blanking)  by  shifting  the  phase  of  the  sweep  until 
the  starting  time  of  horizontal  blanking  and  the  starting  time  of  the  pulse  in  question  ap- 
pear symmetrically  disposed  about  the  center  line  of  the  horizontal  sweep.  C  and  D  may 
then  be  scaled  off  and  the  relative  delay  in  percentage  of  a  horizontal  scanning  period  read 
from  the  chart.  The  delay  time  of  a  network  may  be  similarly  measured  by  comparing 
the  difference  in  position  on  the  sweep  of  an  input  and  output  test  pulse. 

A  variation  of  the  sine-wave  method  uses  an  accurate  phase  shifter  calibrated  in  per- 
centage of  a  scanning  period.  This  does  away  with  chart  and  millimeter  scales,  and  the 
linearity  of  horizontal  sweep  of  the  oscilloscope  is  no  longer  a  factor  as  slopes  and  widths 
may  be  read  off  by  shifting  the  wave  known  amounts  with  respect  to  a  fine  vertical  line 
drawn  down  the  center  of  the  oscilloscope  screen. 

PULSE  CROSS  METHOD.  The  number  of  vertical  synchronizing  signal  sections, 
the  number  of  equalizing  impulses  before  and  after  the  vertical  synchronizing  impulse, 
and  the  approximate  widths  of  the  synchronizing  and  blanking  signals  may  be  determined 
by  the  pulse  cross  method  (reference  35).  These  determinations  are  accomplished  by 
locking  in  the  picture  monitor  with  horizontal  and  vertical  pulses  which  have  been  delayed 
half  a  period  and  reversing  the  polarity  of  the  composite  video  input  signal.  The  pulse 
and  blanking  signals  then  appear  as  a  white  cross  in  the  center  of  the  picture  tube  screen. 
The  items  mentioned  above  may  then  be  determined  by  measurement  and  observation  of 
this  pattern.  The  vertical  deflection  amplitude  should  be  expanded  by  about  3  :  1  while 
making  this  test.  A  simple  switching  arrangement  can  be  arranged  to  shift  the  picture 
monitor  from  normal  to  pulse  cross  operation  rapidly  during  operation. 

19.  OVERALL  VIDEO  SYSTEM  RESPONSE 

In  the  usual  television  plant  the  picture  signals  pass  through  a  relatively  large  number  of 
amplifier  stages  in  cascade  in  traveling  from  the  camera  to  the  transmitter.  The  transient 
response  of  the  overall  system  must  therefore  be  given  careful  consideration.  A  small 
phase  or  amplitude  distortion  in  each  individual  stage  has  a  cumulative  effect  when  a 
large  number  of  stages  is  operated  in  cascade  (reference  39).  The  overall  effect  of  such 
distortion  when  not  compensated  is  to  cause  transients  of  an  oscillatory  nature  to  occur 
whenever  the  scanning  spot  encounters  an  abrupt  change  in  scene  brightness. 

Mathematical  analysis  of  the  overall  transient  characteristics  of  a  practical  system  is 
not  only  difficult  but  is  complicated  by  the  fact  that  all  stages  will  not  ordinarily  use 
identical  forms  of  high-frequency  compensation  or  peaking.  A  practical  engineering  ap- 
proach to  the  problem  which  yields  a  satisfactory  solution  is  the  following:  (a)  When  laying 
out  the  plant,  an  accurate  estimate  can  be  made  of  the  number  of  stages  likely  to  be  con- 
nected in  cascade.  This  estimate  can  be  used  in  conjunction  with  published  data  (refer- 
ence 39)  to  decide  on  reasonable  values  of  design  parameters  for  high-frequency  com- 
pensation of  individual  stages.  (6)  The  design  parameters  for  high-frequency  video  com- 
pensation having  been  chosen,  the  time  delay  distortion,  due  to  a  number  of  stages  in  cas- 
cade, can  be  calculated.  If  the  difference  in  transmission  time  between  medium  fre- 
quencies (100  to  200  kc)  and  high  frequencies  (5  Me)  for  the  number  of  stages  in  cascade 
begins  to  approach  an  appreciable  fraction  of  the  time  of  one  picture  element,  then  it  is 
advisable  either  to  choose  other  design  parameters  giving  smaller  time  delay  distortion 
per  stage  or  to  employ  a  properly  designed  phase  compensation  network,  (c)  It  is  ad- 
visable to  choose  video  amplifier  design  parameters  such  that  the  overall  video  amplitude 
response  will  be  uniform  within  ±1  db.  This  can  be  accomplished  by  choosing  a  top  video 
frequency  (for  design  purposes)  somewhat  higher  than  the  nominal  top  frequency  handled 
by  the  transmitter.  Thus,  the  studio  equipment  amplifiers  might  be  designed  and  com- 


20-36 


TELEVISION 


pensated  for  a  top  frequency  of  6  Me  to  assure  uniform  overall  amplitude  characteristics 
up  to  the  4.5-Mc  nominal  top  frequency  dictated  by  present  standards.  When  this  is 
done,  however,  care  should  be  taken  that  there  are  no  appreciable  peaks  in  the  amplitude 
response  beyond  4.5  Me.  A  gradual  decay  in  amplitude  response  beyond  4.5  Me  rather 
than  an  abrupt  change  is  desirable,  (d)  The  low-frequency  compensation  of  the  video 
system  (60  cycles  to  100  kc)  should  follow  conventional  principles,  considering  the  number 
of  amplifier  stages  in  cascade,  (e)  Having  observed  the  above  design  precautions  and  pro- 
vided a  system  having  uniform  amplitude  response  (within  ±1  db)  from  60  cycles  to  4.5 
or  more  megacycles,  the  overall  transient  characteristics  of  the  system  should  be  investi- 
gated experimentally  by  square-wave  techniques.  The  low-frequency  transient  response 
may  be  investigated  by  means  of  60-cycle  square  waves.  The  high-frequency  transient 
response  may  be  investigated  using  steep-sided  square  waves  of  a  fundamental  frequency 
of  100  kc  (reference  42) .  The  100-kc  square  wave  should  be  of  sufficient  steepness  to  repre- 
sent harmonics  of  100  kc  beyond  the  cut-off  frequency  of  the  system.  The  equivalent 
phase  and  amplitude  characteristics  of  the  system  may  be  obtained  from  the  square-wave 
response  wave  shape  either  by  mathematical  analysis  or  by  means  of  special  charts  (refer- 
ence 40) .  Appropriate  phase  and  amplitude  correction  networks  may  be  designed  on  the 
basis  of  square-wave  test  data. 

Standards  should  be  set  as  to  the  maximum  allowable  amount  of  overshoot  and  follow- 
ing transients  that  can  be  tolerated  in  the  square-wave  response  characteristics  of  a  tele- 
vision broadcast  system.  In  any  specific  case,  however,  one  can  form  an  opinion  as  to 
whether  objectionable  transient  effects  exist  by  observing  the  television  image,  preferably 
an  image  of  a  resolution  test  pattern  having  a  number  of  sharply  defined  boundaries  of 
high  difference  in  contrast.  An  overshoot  or  oscillatory  transient  differing  by  2  per  cent 
from  the  final  steady-state  value  of  a  square  wave  will  normally  be  noticeable  in  the  image. 


20.  TELEVISION  FIELD  PICK-UP  EQUIPMENT 

From  the  electrical  standpoint,  television  field  pick-up  equipment  is  quite  similar  to 
studio  equipment,  the  major  circuit  functions  being  identical.  Mechanically,  however, 
the  equipment  is  usually  segregated  into  small,  light-weight  units  of  suitcase  style  so  that 
it  may  be  readily  transported  (reference  44) . 

A  block  diagram  of  a  typical  complement  of  field  pick-up  equipment  for  two  cameras  is 
given  in  Fig.  10. 

-115  v  60  co  a-c 


Sync. 

115  v_^ 

generator  unit 

60  w  a-c 

and 

power  supply 

:    I 

:    i 

;     i 

;    i 

Pulse  shape 

115  v_^ 
60  w  a-c 

and  delay  unit 
and 
power  supply 

•VSync.  signals 
-^.Driving  pulses 
—>- Video  signal 
Composite  signal 


Camera  control 
unit  1  with 
picture  and 
CRO  monitors, 

Camera  1 



Camera  control 
unit  2  with 
picture  and 
CRO  monitors 

Camera  2 

—  - 

-jfc 

t 

Regulated 
power  supply 

Master  switehlnff 

115v_ 
60  cu  a-c 


115  v_ 
60  w  a-c 


unit  and 
video  line  amp. 


^Reg,  line  out 
i««  »>>-Emg.  line  out 
^•Mon.  out 


Regulated 
power  supply 

1 

1 

Radio  relay  rnoru  In 


FIG.  10.     Block  Diagram  of  a  Typical  Complement  of  Portable  Field  Equipment  for  Two-came: 

Operation 


RELAY  OF  TELEVISION  SIGNAL 


20-37 


FIG.  11.     Typical  Construction  of  a  Coaxial  Cable  Using  a 
Minimum  Amount  of  Solid  Dielectric 


21.  RELAY  OF  TELEVISION  SIGNAL 

Television  network  facilities  may  be  classified  as  "intercity"  and  "local."  Under  the 
first  category  are  included  long-distance  coaxial  carrier  cables  and  radio  relay  multilink 
circuits.  The  latter  class  includes 
existing  telephone  pairs  and  special 
shielded  television  pairs,  video  co- 
axial cables,  and  short-haul  micro-  External  shield 
wave  radio  pick-up  links. 

INTERCITY  TELEVISION  FA- 
CILITIES. Coaxial  cables  (refer- 
ences 45  and  46) .  The  long-distance 
coaxial  cables  consist  of  several  co- 
axials  enclosed  in  a  lead  sheath 
along  with  ordinary  paper-insu- 
lated telephone  pairs.  A  3/s-in. 
longitudinal  seam  coaxial  is  com- 
mon, although  0.27-in.  coaxials 
have  also  been  used.  Figure  11 
pictures  a  coaxial  unit  open  at  the 
end  to  show  the  construction.  See 
also  Section  10. 

Figure  12  shows  the  characteristic  impedance,  the  phase  delay,  and  the  attenuation  of 
3/s-in.  coaxials.  The  useful  frequency  range  for  television  purposes  on  such  a  system  de- 
pends upon  the  type  and  spacing  of 
the  repeaters.  The  present  coaxial 
system  employs  repeaters  at  approxi- 
mately 8-mile  intervals  and  transmits 
television  with  a  311-kc  carrier  up  to 
about  3.1  Me,  giving  a  2.8-Mc  video 
band.  These  lines  must  be  carefully 
equalized  for  gain  and  delay  character- 
istics. Figure  13  shows  overall  char- 
acteristics obtained  (1946)  on  the  New 
York- Washington  coaxial  system. 

Transmission  of  television  signals 
over  local  video  facilities  is  direct;  i.e., 
the  entire  band  of  frequencies  pro- 
duced by  the  television  camera  from 
a  few  cycles  per  second  to  high  fre- 
quencies is  transmitted  directly  over 
the  line.  On  long-distance  coaxial 
facilities,  however,  a  carrier  method  of 
transmission  is  used  to  avoid  the 
effects  of  low-frequency  interference. 
By  using  a  relatively  low  carrier  fre- 
quency and  vestigial  sideband  trans- 
mission, the  band-width  requirements 
of  the  repeaters  are  not  materially  in- 
creased over  a  noncarrier  system. 
Figure  14  illustrates  a  typical  carrier 
transmission  system  for  a  2.8-Mc  pass 
band  in  which  a  dual  modulation  and 
demodulation  process  is  utilized. 

Only  the  frequencies  above  about 
200  kc  are  used  to  transmit  the  tele- 
vision signal  over  the  line,  although 
the  space  below  200  kc  may  be 
used  for  the  accompanying  sound- 

program  channel.      The  main  reason 

5000  for  not   using  very  low   frequencies 

Frequency  -  kc  per  second  is   that   such   coaxial   systems   would 

FIG.  12.     Transmission  Characteristics  of  a  Coaxial  Cable   |?e    noisy    and    difficult    to   equalize 
Constructed  as  in  Fig.  11  (outside  diameter,  3/8inch)       In  the  frequency  range  used,  coaxial 


78 
77 

76 

75 
74 

s, 

Z  =  F 

?+^ 

"*•>• 

v^ 

"^ 

•*-^. 

"--» 

-•—  ^. 

/ 

* 

Te 

TIP 

,  s 

=  5! 

1 

/ 

/ 

/ 

' 

/ 

/ 

'' 

^ 

S 

' 

-^ 

*^^ 

50  70  100     200   500  1000  2000    50( 

20-38 


TELEVISION 


systems  are  extremely  quiet  and  relatively  easy  to  equalize.  For  very  long  distances,  the 
cost  and  complexity  of  the  terminal  apparatus  are  small  compared  to  those  of  the  rest  of 
the  circuit,  but  for  short  video  facilities  they  might  be  objectionable. 


+6 


42 

% 
•      o 

O  -2 


-6 
-8 


+10 


I-1S 

'•-IS 


'has1  s  devatiqn  frcjm'll(ie6rl: 


;  deviation  froi 


400         800 


1200       1600       2000      2400       2800       3200 
Frequency  •  kc 


.     FIG.  13.     Overall  Characteristics  of  the  New  York-Washington  Coaxial  Circuit  (1946) 

Microwave  Radio  Relay  (reference  47)  .  Multilink  radio  relay  systems  also  provide  an 
important  network  facility  for  the  transmission  of  television  programs.  No  general  rules 
can  be  given  governing  the  choice  between  coaxial  cable  and  radio  relay,  however,  as  each 
specific  installation  must  be  carefully  studied  from  a  number  of  standpoints.  Among 
the  factors  governing  such  a  choice,  economic  considerations  will  usually  be  paramount. 
Such  questions  as  relative  operating  and  maintenance  costs  and  network  reliability  must 


Transmitting  terminal 
First  modulation 


Video  signal  band 


Lower  sidebar^ 
(transmit! 


Vestigial  sideband    % 
(partially  transmitted) 


UPP 
S 


'estigial  sideband 

Lower  sideband 
Transmitted  to  line 


4.945 


7.945 
First  carrier 


er  sideband*) 
uppressed-H 

1 


Second  modulation 


10.945 

Upper  sideband 

13.201  to  16.201 

(suppressed) 


8.256 
Second  carrier 


Receiving  terminal 
First  demodulation 


: Upper  sideband J 
(Suppressed) 
2.8| 


45678 
Frequency  In  megacycles  per  second 


11 


12 


FIG.  14.     Diagram  of  a  Typical  Dual  Modulation  and  Demodulation  Process  Used  for  Transmission 
of  a  2.8  Megacycle  Video  Band  over  a  Long  Coaxial  Cable 

also  be  considered  for  each  installation:  Figure  15  indicates  the  repeater  gains  required 
for  both  methods  of  transmission  assuming  a  useful  band  width  of  5  Me.  It  will  be  noted 
that,  except  for  extremely  large-diameter  coaxial  cable,  fewer  repeaters  are  necessary  for 
the  radio  relay  system.  This  figure  also  illustrates  the  reduction  in  repeater  gain  which 
may  be  effected  by  employing  the  higher  frequencies  in  the  microwave  region  when  a  given 
size  antenna  reflector  is  used. 


RELAY  OF  TELEVISION  SIGNAL 


20-39 


160 


LOCAL  TELEVISION  CIRCUIT  FACILITIES.  Various  local  television  transmission 
facilities  may  include  ordinary  telephone  pairs,  specially  designed  shielded  pairs,  coaxial 
cables,  or  microwave  radio  sys- 
tems. Intracity  wire  transmis- 
sion of  television  is  accom- 
plished at  video  frequencies 
thus  avoiding  the  use  of  carrier 
terminals  for  the  short  dis- 
tances encountered. 

Telephone  Pairs  (references 
45  and  46).  Typical  transmis- 
sion characteristics  of  tele- 
phone pairs  are  shown  in  Fig. 
16.  Because  of  the  large  at- 
tenuation of  the  higher  video 
frequencies,  it  is  necessary  to 
install  suitable  amplifiers  at 
spacings  of  an  average  length 
of  1  mile,  though  this  interval 
may  be  increased  somewhat 
when  the  larger  gages  are  em- 
ployed. 

It  is  necessary  to  equalize  the 
attenuation  and  phase  charac- 
teristic vs.  frequency  of  the 
telephone  pairs  over  the  video 
band  of  frequencies.  This  is 
accomplished  by  means  of  vari- 
able equalizers  associated  with 
each  video  amplifier.  Pre-em- 
phasis  of  the  level  of  the  higher 
video  frequencies  is  usually  em- 
ployed in  these  transmission 
circuits  to  obtain  an  improved 


15  20 

Repeater  spacing-miles 


signal-to-noise  ratio. 


Comparative  Repeater  Spacings  and  Gains  for  Coaxial 
and  Microwave  Circuits 


FIG.  15. 

Other  factors  to  be  consid- 
ered in  the  use  of  telephone  pairs  for  video  transmission  are  the  selection  of  suitable 
pairs  within  the  cables  and  the  removal  of  bridged  taps  on  the  pair  selected.  The  selection 
of  pairs  within  a  chosen  cable  sheath  is  made  with  a  view  toward  reducing  interference 


20  40      60  80100        200         400    600    1000       2000       4000- 

Frequency  In  kilocycles  per  second 

FIG.  16.     Insertion  Loss  of  Various  Types  of  Telephone  Pairs  for  Video  Frequencies 

from  adjacent  circuits  and  to  avoid  cross-talk  coupling  around  the  video  repeater.  An 
bridged  taps  are  removed  to  assure  minimum  imped-knce  irregularities  due  to  these  lumped 
constants  along  the  lines. 


20-40 


TELEVISION 


Shielded  Pairs.  Where  it  is  desirable  to  extend  the  length  of  the  wire  circuit  between 
video  repeaters,  the  use  of  special  shielded  pair  is  indicated.  An  opened  section  of  such  a 
cable  is  shown  in  Fig.  17.  The  transmission  characteristics  of  such  a  shielded  pair  are 


FIG.  17.     Low  Loss  Balanced  Shielded  Video  Transmission  Line 

given  in  Fig.  18.     With  this  pair  it  is  possible  to  extend  the  video  repeater  spacing  to  ap- 
proximately 3.5  miles. 

This  type  of  cable  consists  of  a  pair  of  No.  16  gage  wires  insulated  and  spaced  by  means 
of  polyethylene  strings.  This  core  is  then  enclosed  within  a  cylindrical  shield  formed  with 
metal  tapes.  A  number  of  these  units  may  be  enclosed  in  a  lead  sheath,  or  the  shielded 

pair  unit  may  be  inserted  in  an 
ordinary  telephone  cable  re- 
placing a  certain  number  of  the 
usual  paper-insulated  pairs. 

Coaxial  Cable  (references  45 
and  46) .  Although  of  primary 
importance  for  network  facili- 
ties, coaxial  cable  may  be  used 
for  intracity  circuits  of  a  more 
permanent  nature,  such  as  stu- 
dio-transmitter link  service. 
For  short  distances,  transmis- 
sion of  the  video  signals  with- 
out utilizing  a  carrier  system  is 
practicable.  The  inherent  un- 
balanced properties  of  the  co- 
axial cable,  however,  usually 
re  that  special  low-fre- 
quency balancing  circuits  be 
employed.  A  typical  "hum" 
balancing  circuit  is  shown  in 


16 
14 
12 
10 
3 
6 
4 
2 
0 

/ 

/ 

/ 

/ 

/ 

/ 

s 

' 

' 

^ 

^ 

^~ 

.-.-• 

.^-^ 

50    70   100          200               500         1000       2000             5QC 
Frequency-kc  per  second 

FIG.  18.     Attenuation  Characteristics  of  the  Line  Illustrated  in 
Fig.  17 


Fig.  19.  Included  in  this  figure  is  a  simulated  generator  of  low-frequency  interference. 
It  will  be  noted  that  noise  current  components  flow  in  opposite  directions  through  the  cable 
terminating  resistor  RI  and  the  hum  balancing  potentiometer  R2.  By  adjustment  of  the 
balancing  potentiometer,  a  condition  can  be  found  such  that  equal  and  opposite  noise 
voltages  are  developed  between  points  a-b  and  points  b-c.  The  noise  is  thereby  reduced 


--j"  conductor 
InsuJated  from  shield 


r—ty—— ------ 

5  PCoaxiaJ  cable  grounded  at 

r^  j         Input  terminal  only 


'^Balancing  potentiometer 
Jd 


•=•  Local  ground  at  output 
Sunulated  Interference  generator         termina'  °f  Cab'e 

FIG.  19.     Hum  Balancing  System  for  a  Coaxial  Cable  Having  an  External  Lead  Sheath  Insulated 
from  the  Copper  Outer  Conductor 

at  the  input  of  the  amplifier  so  that  the  signal  voltages  alone  are  applied  between  the 
grid  and  cathode  of  V\. 

Equalization  of  transmission  of  coaxial  cables  is  usually  accomplished  by  the  use  of  an 
equalizing  network  inserted  between  the  cable  and  the  input  of  the  receiving  video  amplifier. 


TRANSMITTER  PLANT  TERMINAL  EQUIPMENT       20-41 

In  certain^  applications  where  coaxial  and  balanced  transmission  facilities  are  to  be  con- 
nected, special  amplifier  circuits  or  wide-band  video  repeating  coils  are  employed  to  make 
the  transition. 

Microwave  Radio  Relay.  Microwave  radio  relay  is  also  an  important  local  facility 
for  video  transmission.  Many  television  field  programs,  such  as  parades  and  special 
events  broadcasts,  are  not  repeated  at  frequent  enough  intervals  from  a  given  location  to 
justify  the  expense  of  wire  facility  installations.  Furthermore,  programs  originating  at 
distances  greater  than  20  miles  from  the  studio  plant  are  usually  more  economically  handled 
by  radio  relay.  Proper  evaluation  of  these  and  related  factors  is  necessary  to  determine 
the  choice  between  radio  relay  or  wire  facilities.  Radio  relay  may  also  be  employed  for 
studio-transmitter  link  service. 

The  power  requirements  for  a  microwave  relay  system  may  be  approximated  by  the 
following  formula.  Although  accurate  only  for  the  free-space  propagation  condition, 
application  of  this  formula  in  practice  where  line  of  sight  exists  will  yield  results  of  suf- 
ficient accuracy  to  be  useful.  It  should  be  noted  that  the  maximum  total  received  power 
due  to  ground  reinforcement  can  approach,  as  a  limit,  four  times  the  received  power  for 
free-space  propagation  as  obtained  from  the  formula. 

The  free-space  transmission  formula  is  (reference  48)  : 


-r-o- 

where  Pt  =  power  fed  to  transmitting  antenna  at  input  terminals       1  « 

pr  =  power  available  at  output  terminals  of  receiving  antenna/  um 

Ar  —  effective  area  of  receiving  antenna 


At  =  effective  area  of  transmitting  antenna 
d  —  distance  between  antennas 


Same  units 


X  ss  wavelength 

The  power  necessary  at  the  receiving  antenna  output  terminals  depends,  among  other 
factors,  upon  the  signal-to-noise  ratio  requirements  of  the  relay  system.  The  noise  level 
due  to  thermal  agitation  at  20  deg  cent  may  be  computed  from  the  following  expression 
(reference  47) : 

Pn  =  (0.8  X  10~20)  (5)  (6) 

where  Pn  =  noise  power  in  watts  due  to  thermal  agitation. 

B  =*  twice  the  highest  modulation  frequency  in  cycles  per  second. 

In  practice,  the  noise  level  due  to  all  equipment  causes  will  usually  be  between  10  and  15 
db  above  thermal. 

The  effective  area  of  an  antenna  is  directly  proportional  to  the  power  gain.  The  follow- 
ing tabulations  indicate  the  effective  areas  of  several  typical  antennas. 

ANTENNA  EFFECTIVE  AREA 

Isotrot)ic  radiator \z/4-7r 

Half-wave  dipole 0. 1305X2 

Parabolic  reflector Two-thirds  of  the  projected 

area  of  the  paraboloid 

Several  factors  affect  the  effective  area  of  a  paraboloid,  the  most  important  being  the 
efficiency  of  excitation.  For  example,  the  effective  area  is  reduced  to  approximately 
three-eighths  of  the  projected  area  when  only  half  of  the  exciting  antenna  energy  is  directed 
toward  the  reflector.  If  transmission  lines  or  wave  guides  are  used  in  the  antenna  system, 
the  attenuation  due  to  these  components  should  be  taken  into  consideration  when  apply- 
ing the  free-space-transmission  formula. 

22.  TRANSMITTER  PLANT  TERMINAL  EQUIPMENT 

Terminal  equipment  located  at  the  television  transmitting  plant  performs  the  function 
of  raising  the  signal  level  delivered  by  the  program  source  to  that  required  by  the  trans- 
mitter and  provides  the  necessary  picture  and  wave-form  monitoring  facilities.  The  equip- 
ment is  usually  installed  in  a  shielded  room;  it  may  consist  of  the  following  units: 

A.  An  amplifier  with  means  of  controlling  the  composite  signal  amplitude. 

B.  An  amplifier  with  independent  amplitude  control  of  the  synchronizing  and  picture 
portions  of  the  composite  video  signal. 

C.  Video  switching  system  to  select  one  of  several  sources  of  signal. 

D.  A  line  amplifier  of  sufficient  output  to  meet  the  input  level  requirements  of  the  video 
section  of  the  transmitter. 


20-42 


TELEVISION 


B.  Picture  and  wave-form  monitors. 

F.  Monitor  switching  system  to  select  circuits  to  be  monitored. 

Figure  20  shows  a  simplified  block  diagram  of  the  visual  portion  of  a  typical  television 
transmitter  plant.  Equipment  required  for  the  television  sound  channel  follows  standard 
frequency-modulation  broadcasting  practice. 

VISUAL  CARRIER  FREQUENCY  GENERATION.  The  r-f  carrier  signal  for  a  tele- 
vision transmitter  is  generally  developed  by  conventional  methods.  The  primary  source 
of  radio-frequency  energy  is  usually  a  highly  stabilized  quartz-crystal  oscillator  operated 
at  a  relatively  low  frequency.  This  low-frequency,  low-power  signal  is  multiplied  and 


From  relay 
facility 


FIG.  20.    Picture  Transmitter  Block  Diagram  Showing  Video  Input  Equipment 

amplified  to  the  frequency  and  power  level  required  at  the  modulated  amplifier  stage  of 
the  transmitter. 

MODULATION  METHOD.  A  few  of  the  many  possible  methods  of  modulating  the 
visual  carrier  (reference  49)  are  illustrated  in  Fig,  21.  Of  these,  grid-bias  modulation  is 
almost  universally  used. 

Modulation  may  be  either  at  low  or  high  r-f  level.  At  low  level  the  grid-bias-modulated 
r-f  amplifier  is  followed  by  one  or  more  class  B  linear  r-f  amplifier  stages  having  the  re- 
quired band-pass  characteristics. 

MODULATED  AMPLIFIER.  The  plate  and  grid  (if  grid-bias  modulation  is  used)  tank 
circuits  of  the  modulated  amplifier  as  well  as  all  succeeding  r-f  stages  must  be  capable  of 
passing  the  generated  sideband  power  without  excessive  amplitude  or  phase  distortion. 
This  requires  tank  circuits  of  relatively  low  impedance  resulting  in  rather  poor  operating 
efficiencies  as  compared  to  sound  transmitters. 

NEUTRALIZATION.  The  band-pass  characteristic  of  a  television  transmitter  using 
triode  tubes  depends  not  only  upon  the  circuit  elements  but  also  upon  the  effectiveness  of 
neutralization.  At  low  frequencies,  where  lead  inductances  may  be  neglected,  a  simple 
capacitance  bridge  adequately  represents  the  neutralizing  circuits.  At  higher  frequencies, 
however,  where  lead  inductances  become  appreciable,  additional  compensation  is  usually 
necessary,  especially  for  wide-band  operation.  Figure  22  indicates  the  stray  inductances 
often  encountered  and  methods  of  compensation  (reference  50) .  Stray  or  undesired  cou- 
pling between  input  and  output  circuits  not  only  disturbs  neutralization  but  also  affects 
the  band-pass  characteristics  of  the  amplifier. 


TRANSMITTER  PLANT  TERMINAL  EQUIPMENT       20-43 

D-C  TRANSMISSION.  According  to  1946  standards,  the  tips  of  the  synchronizing 
signals  correspond  to  maximum  carrier  envelope  amplitude,  and  this  is  held  as  nearly  con- 
stant as  possible  during  a  given  transmission.  Black  level,  which  corresponds  to  the  base 
of  the  synchronizing  pulses,  is  maintained  at  a  fixed  percentage  of  the  maximum  carrier 
envelope  amplitude  within  narrow  tolerance  independent  of  the  values  of  light  and  shade 
in  the  picture  transmitted.  In  order  to  achieve  this  result  one  or  more  of  the  various  forms 
of  "d-c  restoration"  circuits  are  used  in  the  modulator  stages  of  the  transmitter.  (See 
article  7.) 


Modulator 


Grid-modulated  r-f  amplifier 


Grid-bias  modulation 

Plate-modulated  r-f  amplifier 
Modulator 


Plate  modulation 


R-f  amplifier  L   90°  at 


4M, 

^J^F^ 

to 

^_ 

output 

O 

t 

T 

<j> 

-,- 

L  

Load  Impedance  modulation 
FIG.  21.     Video  Modulation  Methods  for  Amplitude  Modulation 

VESTIGIAL  SIDEBAND  TRANSMISSION.  A  vestigial  sideband  system  of  television 
transmission  (reference  51)  is  standard  for  commercial  television  broadcasting.  The 
higher-frequency  sideband  components  up  to  but  not  in  excess  of  carrier  plus  4.5  Me  and 
the  lower-frequency  sideband  components  down  to  carrier  frequency  —0.75  Me  are  trans- 
mitted. The  remainder  of  the  low  sideband  energy  must  be  attenuated  as  rapidly  as 
possible  and  must  reach  and  retain  a  low  order  of  magnitude  at  frequencies  lower  than 
carrier  frequency  —1.25  Me.  Curve  C  of  Fig.  20,  article  9 r  illustrates  the  standard 
vestigial  sideband  transmission  characteristic. 

Transmitters  which  are  modulated  at  high  power  level  must  be  followed  by  a  vestigial 
sideband  filter  which  absorbs  the  developed  but  undesired  sideband  energy  (reference 
52) .  Vestigial  sideband  filters  employ  various  configurations  of  elements  and  are  usually 
constructed  in  the  form  of  sections  of  concentric  transmission  lines  of  suitable  lengths, 
diameters,  and  diameter  ratios.  A  single  section  of  one  type  of  vestigial  sideband  filter 
is  shown  schematically  in  Fig.  23.  Generally,  two  or  more  sections  of  this  type  of  filter 
are  used.  In  addition,  "notching"  filters  are  often  required  to  provide  additional  at- 
tenuation at  the  low-frequency  edge  of  the  assigned  channel. 

Transmitters  which  are  modulated  at  low  power  level  may  achieve  the  required  trans- 
mission characteristics  by  proper  design  and  adjustment  of  the  interstage  coupling  net- 
works of  the  following  linear  amplifiers.  In  this  case,  a  less  pretentious  vestigial  sideband 
filter  may  be  necessary  at  the  output  of  this  type  of  transmitter  in  order  to  achieve  the 
required  characteristic. 


20-44 


TELEVISION 


{    RADIO-FREQUENCY  M ONITORING.    Since  the  output  of  a  vestigial  sideband  trans- 
mission system  is  viewed  on  receivers  having  specified  band-pass  characteristics,  a  mon- 


Brldge  neutralization 
Indicating  disturbing 
self-inductances 


CN 


1  —  ir- 

G,c=£ 

—  ^ 

f   ,  T 

f  +i 

T  f-wv 

IGni  1—  i 

Basic  circuit 


CM 


Bridge  neutralization 
Indicating  compensation 
of  disturbing  Inductances 
by  capacitors  Cr  Ca/  and  C8 


FIG.  22.     Neutralization  Method  to  Compensate  for  Stray  Inductance 

itor  should  be  provided  which  not  only  conforms  to  the  standard  receiver  characteristic 

but  also  yields  a  signal  that  is  a  true  sample  of  the  radiated  energy  from  the  transmitter. 

The  aural  transmitter  may  be  monitored  with  equipment  similar  to  that  developed  for 

the  frequency-modulation  broadcast  serv- 
ice. 

MODULATION  MEASUREMENT. 
The  modulation  of  the  television  trans- 
mitter may  be  measured  in  various  ways, 
but  the  methods  that  take  advantage  of 
the  fact  that  a  television  transmitter  is 
operated  at  a  constant  peak  carrier  level 
have  been  found  most  satisfactory  in 
practice. 

Since  the  tips  of  the  synchronizing 
signals  represent  100  per  cent  modula- 
tion, one  relatively  simple  method  is  to 
observe  the  carrier  envelope  pattern  at 
radio  frequency  on  an  oscilloscope  as 
shown  in  Fig.  24.  At  the  high  carrier 


FIG.  23.     Single  Section  Vestigial  Sideband  Filter 


frequencies  involved,  however,  it  is  sometimes  difficult  to  insure  that  the  cathode-ray 
pattern  is  a  true  representation  of  the  developed  carrier  envelope  amplitude. 


TRANSMITTER   PLANT  TERMINAL  EQUIPMENT      20-45 

A  method  which  avoids  dealing  with  the  radio  frequency  directly  utilizes  the  output  of  a 
linear  rectifier  applied  to  the  vertical  plates  of  the  oscilloscope,  normal  sawtooth  sweep  of 
a  convenient  frequency  being  used  for  horizontal  deflection.  A  contactor  is  provided  to 
short-circuit  the  output  of  the  diode  periodically.  This  provides  a  reference  level  cor- 
responding to  complete  modulation  in  the  white  direction,  or  zero  carrier  envelope  am- 
plitude. The  modulation  percentage  may  be  scaled  off  the  cathode-ray  screen  as  in- 
dicated in  Fig.  24  which  shows  the  appearance  of  the  pattern  using  high  frequency  sweep. 
This  simple  method  is  quite  accurate  and  may  be  utilized  periodically  as  a  visual  operating 
check  of  transmitter  modulation  during  a  program  (reference  53) . 


Carrier  envelope 


H 

orlzontal  sync 
r—  pulse 

/  j 

I 

110 

**, 

L 

*T 

\l  I 

r, 

2 

Time  —  ^ 

Horizontal  sync 
1       T.X  pulse 

Zero  reference 
(Contactor  closed) 

<{ 


FIG.  24.     Video  Modulation  Measurement,  (a)   Oscilloscope  presentation  of  video  modulated  RF 
envelope  (HF  sweep).     (6)  Oscilloscope  presentation  of  output  of  linear  detector  with  shorting  con- 
tactor (HF  sweep). 

MEASUREMENT  OF  R-F  OUTPUT  POWER.  Television  transmitters  are  rated  in 
peak  power  output,  i.e.,  the  power  output  level  attained  during  the  synchronizing  pulse 
portion  of  the  transmitted  signals. 

The  following  methods  of  measuring  power  output  assume  that  the  transmitter  power 
output  can  be  held  at  the  operating  peak  level.  For  transmitters  that  cannot  be  held  at 
peak  output  for  measurement  purposes,  other  methods  have  to  be  used  or  a  reliable  cor- 
rection factor  must  be  applied. 

The  transmitter  output  power  may  be  determined  by  measuring  the  power  delivered 
to  a  water-cooled  resistance  load  with  circuit  adjustments  capable  of  transmitting  a  good 
picture.  The  rate  of  water  flow  and  the  temperature  rise  of  the  water  stream  on  passing 
over  the  resistor  must  be  accurately  measured.  Then  the  power  delivered  is  given  by 

P  =  2QZTF  (7) 

where  P  =  power  delivered  to  load  in  watts. 

T  =  temperature  change  of  water  in  degrees  centigrade. 
F  =  water  flow  in  gallons  per  minute. 

It  is  often  desirable  to  know  the  power  delivered  to  the  actual  operating  load.  Since 
most  practical  transmission  line  installations  are  not  perfectly  matched  to  the  radiator, 
a  finite  reflection  occurs  on  the  line.  It  is,  therefore,  necessary  to  determine  the  average 
value  of  transmission  line  voltage.  One  satisfactory  method  utilizes  a  slotted  section  of 
transmission  line  and  a  calibrated  vacuum-tube  voltmeter  for  determining  the  maximum 
and  minimum  values  of  the  transmission  line  voltage. 

The  power  delivered  to  the  transmission  line  load,  neglecting  line  attenuation,  may  be 


20-46  TELEVISION 

calculated  by  means  of  the  following  formula,  which  is  accurate  to  better  than  1  per  cent 
if  the  voltage  standing  wave  ratio  is  between  0.8  and  unity. 

ln) 


where  Po  =  power  delivered  to  load  in  watts. 

jEWx  =  rms  value  of  voltage  maximum  in  volts. 
•#mfn  ~  rms  value  of  voltage  minimum  in  volts. 

ZQ  =  characteristic  impedance  of  transmission,  line  in  ohms. 

Alternatively,  the  output  power  may  be  determined  by  permanently  locating  two  cal- 
ibrated vacuum-tube  voltmeters  precisely  one-quarter  wavelength  (electrical)  apart  and 
substituting  the  voltage  indicated  by  these  meters  for  jE/max  and  j&min  in  the  above  formula. 

TRANSMISSION  LINE.  The  transmission  line  system  between  the  transmitter  and 
the  antenna  must  be  well  matched  over  the  band  of  frequencies  which  includes  the  carrier 
and  the  sideband  frequencies  of  appreciable  magnitude.  Multiple  images  may  appear 
in  the  radiated  signal  of  a  poorly  matched  transmission  system.  Satisfactory  results  are 
usually  produced  when  a  standing  wave  ratio  between  0.9  and  1.0  over  the  required  band 
of  frequencies  exists. 

ANTENNAS.  Commercial  television  broadcasting  antennas  are  required  to  be  hori- 
zontally polarized.  The  directivity  and  radiating  efficiency  of  the  antenna  should  be  sub- 
stantially independent  of  frequency  over  the  desired  transmission  band.  The  input  im- 
pedance of  the  antenna  must  be  substantially  independent  of  frequency  and  must  match 
the  transmission  line  well  enough  to  avoid  the  development  and  transmission  of  multiple 
images  (references  54  and  55)  .  Appreciable  effective  power  gain  may  be  obtained  by  com- 
pressing the  radiated  energy  in  the  vertical  plane. 

PERFORMANCE  MEASUREMENTS.  The  response  of  the  visual  transmitter  from 
input  to  radio  monitor  may  be  measured  using  sinusoidal  modulation.  The  modulating 
frequency  should  be  varied  incrementally  over  the  required  band  while  the  relative  re- 
sponse as  a  function  of  frequency  is  measured  on  a  cathode-ray  oscilloscope  or  vacuum-tube 
voltmeter.  Alternatively,  the  sinusoidal  modulating  signal  may  be  injected  in  the  studio 
equipment  before  the  blanking  signals  are  added,  and  the  relative  response  at  the  trans- 
mitter input  and  at  the  radio  monitor  output  may  be  measured  on  a  cathode-ray  oscillo- 
scope. This  method  is  applicable  to  transmitting  systems  which  use  d-c  restoration  cir- 
cuits, and  the  results  are  representative  of  what  may  be  expected  under  actual  operating 
conditions. 

Means  of  observing  and  recording  the  transient  response  of  a  television  transmission 
system  are  desirable.  If  a  100-kc  square  wave,  having  a  rise  time  which  is  short  compared 
to  the  rise  time  expected  from  the  circuit  under  test,  is  applied  to  the  transmitter  the 
transient  response  may  be  observed  on  a  cathode-ray  oscilloscope  connected  to  a  r-f  mon- 
itor. Some  square-wave  generators  provide  not  only  a  100-kc  square-wave  output  but 
also  a  synchronous  100-kc  sinusoidal  output  and  a  synchronous  10-  or  20-Mc  output. 
The  100-kc  sinusoidal  output  may  be  advantageously  used  for  horizontal  deflection  of  the 
measuring  oscilloscope.  The  20-Mc  output  may  be  used  to  modulate  the  cathode-ray 
beam  in  amplitude.  When  the  cathode-ray-tube  bias  is  properly  adjusted  only  the  pos- 
itive peaks  of  the  20-Mc  modulation  are  visible,  thus  providing  an  accurate  time  base 
which  may  be  used  to  measure  the  change  in  amplitude  of  the  signal  for  each  accurate 
time  interval.  The  realized  rise  time,  as  well  as  the  magnitude  of  overshoots  or  oscil- 
latory transients,  may  thus  be  accurately  determined.  Such  performance  measure- 
ments can  generally  be  correlated  directly  with  the  appearance  of  the  reproduced  television 
image. 

TELEVISION  RECEIVERS 

By  W.  F.  Bailey  and  R.  J.  Brunn 

Receivers  for  television  signals  in  accordance  with  the  present-day  standards  of  the 
Federal  Communications  Commission  are  of  the  superheterodyne  type  and  receive  both 
the  picture  and  sound  transmissions.  A  block  diagram  of  a  typical  television  receiver  is 
shown  in  Fig.  1. 

The  picture  and  sound  carrier  signals  are  received  by  a  single  antenna  and  are  am- 
plified in  a  single  channel.  The  selectivity  of  this  channel  protects  against  image  signal 
and  cross-modulation  interference.  Frequency  conversion  and  some  amplification  at  the 
intermediate  frequencies  are  also  accomplished  in  a  single  channel.  Then  the  picture  and 
sound  signals  are  separated  and  each  is  amplified  sufficiently  for  final  detection.  The 


K-F  CIRCUITS 


20-47 


selectivity  in  each  channel  must  be  adequate  to  keep  the  sound  and  picture  signals  from 
interfering  with  each  other,  and  also  to  attenuate  adjacent  signals  to  a  non-interfering 
Level.  An  f-m  detector  and  an  audio  amplifier  complete  the  sound  channel. 

In  the  picture  channel,  following  the  second  detector,  amplification  occurs  at  video 
frequency.  Either  direct  coupling,  or  d-c  restoration,  or  a  combination,  is  used  to  main- 
tain the  voltage  corresponding  to  black  constant  at  the  picture  tube.  In  first-grade  re- 
ceivers, both  automatic  gain  control  and  noise  limiting  are  provided  in  the  picture  channel. 

Synchronizing  signals  are  extracted  from  the  complete  picture  signal  and  are  separated 
for  the  respective  scanning  oscillators. 

Scanning  generators  produce  sawtooth  waves  at  line  and  field  frequency  of  either  cur- 
rent or  voltage,  depending  on  the  type  of  picture  tube.  Magnetic  deflection  is  commonly 
used  for  best  resolution  since  there  is  less  defocusing  than  there  is  with  electric  deflection. 


Transmission 
line 

rl 

•*jj     Loudspeaker 

Picture 
tube      .f 

|c=5r 

Sound 
amplifier 

» 

F-m 
detector 

- 

A-f 

amplifier 

~  ampler  *  Modu" 

Common  J 
or  •»       |.f 
amplifier  ~| 

1  (            1 
Sound  avc 

1                Local 
]              oscillat 

L 

or 

Picture 
I-f 

amplifier 

* 

Picture 
detector 

* 

Video 
amplifier 

r              n  T              1 
J     Noise    ij      D-C 
"Tf    Hmiter    y    relnserter 

1 

Picture 
automatic 
tain  control 

^ 

1 

"Synchroniz-J 

Line 
scanning 
generator 

* 

]  separator  |1 

Field 
scanning 
generator 

Low 
voltage 
supply 

High 
voltaqe 
supply 

FIG.  1.    Block  Diagram  of  a  Television  Receiver 

The  d-c  accelerating  potential  for  the  picture  tube  is  obtained  by  rectifying  the  power- 
frequency  wave,  an  r-f  sine  wave,  or  the  voltage  impulse  during  the  line  retrace  in  a  mag- 
netic scanning  system. 

23.  ANTENNAS 

The  usual  antenna  for  reception  is  a  half-wave  dipole  in  which  the  radiator  diameter  is 
from  0.2  to  2.5  per  cent  of  the  antenna  length,  to  improve  performance  over  the  frequency 
range.  The  folded  dipole  antenna  (reference  55)  is  also  used  with  a  300-ohm  transmission 
line.  Some  use  has  been  made  of  a  reflector  to  improve  the  directivity,  but,  with  a  simple 
array,  not  much  directive  gain  can  be  obtained  over  the  frequency  range. 

Usually,  a  balanced  line  (reference  57)  of  75  to  300  ohms  impedance  is  used  to  transmit 
the  signal  from  the  antenna  to  the  receiver. 


24.  R-F  CIRCUITS 

REQUIREMENTS.  The  r-f  circuits  of  a  television  receiver  couple  the  signal  from  the 
antenna  transmission  line  to  the  modulator.  The  following  factors  must  be  considered  in 
the  r-f  circuit  design:  (1)  r-f  gain;  (2)  band  width;  (3)  selectivity;  (4)  coupling  to  trans- 
mission line;  (5)  station  selection;  (6)  oscillator  radiation;  (7)  noise  factor. 

R-F  AMPLIFIER.  Normally,  it  is  not  necessary  to  provide  amplification  at  radio 
frequency  because  it  is  easier  to  obtain  the  necessary  amplification  at  intermediate  fre- 
quency. However,  a,n  r-f  stage  is  helpful  in  reducing  oscillator  radiation  (reference  58) 
since  there  may  be  an  attenuation  of  10  to  50  times  for  signal  propagation  in  the  back- 
ward direction  through  the  r-f  stage. 

Most  modulators  have  considerable  noise  (references  59  and  60) .  The  inherent  noise  of 
the  receiver  may  sometimes  be  reduced  by  use  of  an  r~f  stage  (reference  61) .  A  triode  r-f 
amplifier  will  reduce  the  noise  to  the  minimum.  Generally,  a  pentode  r-f  amplifier  will 
provide  no  improvement.  In  order  to  eliminate  the  need  for  neutralization,  the  triode 
is  generally  used  in  a  grounded-grid  circuit,  and  a  tube  having  a  low  plate-to-cathode 
capacitance  is  chosen. 


20-48 


TELEVISION 


ANTENNA  COUPLING.  With  a  low-loss-transmission  line,  it  is  desirable  that  the 
receiver  input  circuit  match  the  line  with  a  low  standing  wave  ratio  to  eliminate  ghosts  in 
the  picture  caused  by  multiple  traversals  of  the  transmission  line  (reference  62).  The 
antenna  cannot  be  expected  to  terminate  the  line  with  a  standing  wave  ratio  lower  than 
about  10  db  in  some  of  the  channels.  Thus  the  reflection  at  the  receiver  must  be  kept  low. 
With  2-db  attenuation  in  the  line  in  one  traversal,  and  an  antenna  termination  with  a  10- 
db  standing  wave  ratio,  it  is  necessary  that  the  receiver  terminate  the  transmission  line 
with  a  standing  wave  ratio  of  0.6  db  so  that  the  signal-to-ghost  ratio  in  the  receiver  be  40 
db. 

A  resistive  element  must  be  present  in  the  receiver  to  achieve  termination  of  the  trans- 
mission line  with  a  low  standing  wave  ratio.  The  input  conductance  of  the  first  tube,  the 


Gang  switch 


FIG.  2.     Typical  Antenna  to  Modulator  Coupling  Circuits 

inherent  losses  in  the  reactive  circuit  elements,  and,  in  some  cases,  a  resistor  added  for 
the  purpose,  constitute  the  source  of  loading  for  the  input  circuit.  The  added  resistor  to 
produce  the  required  loading  increases  the  noise  factor  above  the  minimum  (reference  63). 

To  realize  the  benefit  of  the  balanced  line  in  minimizing  extraneous  pick-up,  a  coupling 
circuit  must  be  used  between  the  transmission  line  and  the  first  tube  which  has  good 
transmission  for  balanced  signals  and  high  attenuation  for  unbalanced  signals.  This  will 
reduce  interference  picked  up  by  the  transmission  line  acting  as  a  single-wire  antenna. 

Transformers  in  which  the  electrostatic  coupling  is  minimized  are  used  to  couple  to  the 
balanced  line.  The  secondary  is  usually  unbalanced  to  deliver  the  signal  to  a  single  grid 
or  cathode.  The  transformer  may  be  coupled  directly  into  a  cathode  with  a  good  im- 
pedance match,  but  no  selectivity. 

SELECTIVITY  REQUIREMENTS.  Because  the  picture  and  sound  carriers  are  at 
opposite  ends  of  a  television  channel,  the  r-f  circuits  should  have  substantially  uniform 


PICTURE  I-F  AMPLIFIER  20-49 

transmission  over  a  band  width  of  about  5  Me.  This  is  required  because  the  sound  car- 
rier, the^main  sideband,  and  the  vestigial  sideband  should  be  amplified  uniformly. 

Sufficient  selectivity  should  be  provided  to  give  an  image  ratio  of  at  least  40  db.  This 
provides  protection  against  image  signals  which  are  as  strong  as  the  desired  signal.  This 
order  of  selectivity  requires  a  minimum  of  two  tuned  circuits. 

STATION  SELECTION.  Station  selection  may  be  accomplished  by  switching,  with 
fixed  or  movable  coils,  or  by  continuously  adjustable  inductors,  using  tuning  cores,  or  by 
varying  the  length  of  the  wire  to  change  the  inductance. 

The  requirements  for  tuning  are:  (1)  station  selection  by  a  single  control;  (2)  reliable 
long-life,  noise-free  operation;  (3)  provision  for  a  number  of  channels  lying  between  4  and 
12;  (4)  resett ability. 

Some  typical  r-f  circuits  between  the  antenna  transmission  line  and  the  modulator  are 
shown  in  Fig.  2  (references  64  and  65) . 

25.  MODULATOR  AND  LOCAL  OSCILLATOR 

MODULATOR.  The  tubes  commonly  used  for  the  modulator  are  the  triode  and 
pentode  types.  Multigrid  converters  are  rarely  used  because  of  their  high  noise  (refer- 
ences 59  and  60) . 

Because  it  has  the  lowest  internal  noise,  the  triode  modulator  is  used  when  the  best 
noise  factor  is  desired.  However,  the  triode  presents  design  problems,  since  both  the  in- 
put and  the  output  impedances  are  functions  of  the  oscillator  excitation,  and  the  grid-to- 
plate  capacitance  makes  the  input  and  output  circuits  interdependent. 

The  pentode  modulator  does  not  produce  as  low  a  noise  factor  because  of  the  partition 
noise.  It  is  less  difficult  to  use  in  the  receiver,  since  there  is  negligible  coupling  between 
the  input  and  output  circuits.  Also,  the  output  impedance  of  a  pentode  modulator  is 
normally  so  high  that  variations  in  it,  caused  by  changes  of  oscillator  excitation,  have  no 
effect  on  the  performance  of  the  i-f  amplifier. 

A  high  transconductance  tube  is  used  to  maintain  high  conversion  gain  and  low  noise. 
It  is  usual  to  bias  the  modulator  by  drawing  grid  current  on  the  local  oscillator  signal. 

LOCAL  OSCILLATOR.  The  local  oscillator  is  usually  a  triode  used  with  either  capaci- 
tive  or  inductive  feedback.  Capacitive  feedback  offers  the  advantages  that  the  inherent 
capacitances  of  the  tube  may  be  used  directly  in  the  oscillator  circuit  to  produce  feedback, 
and  that  the  tuning  coil  has  no  tap. 

LOCAL  OSCILLATOR  DRIFT.  Both  the  picture  and  sound  quality  suffer  if  the  oscil- 
lator varies  from  the  correct  frequency  either  by  oscillator  drift  or  poor  resettability  of 
the  tuning  device.  Frequency  stability  is  of  prime  importance.  A  frequency  shift  of 
±150  kc  is  about  the  upper  limit  that  can  be  tolerated  by  the  picture.  This  drift  will 
produce  approximately  ±2-db  variation  in  the  amplitude  of  low-frequency  video  com- 
ponents relative  to  high-frequency  components.  The  cost  of  the  sound  channel  is  increased 
with  high  shifts.  The  sound-channel  band  width  and  the  linear  portion  of  the  detector 
characteristic  must  be  adequate  to  accommodate  the  drift.  Otherwise  the  sound  i-f  may 
lie  on  the  side  of  the  i-f  amplifier  transmission  characteristic.  This  produces  amplitude 
modulation  of  the  f-m  signal.  Further,  the  performance  of  the  frequency  detector  may 
suffer,  as  the  signal  may  be  near  one  of  the  peaks  and  will  be  on  a  non-linear  part  of  the 
detector  characteristic. 

Receivers  have  frequently  employed  an  oscillator  tuning  adjustment  so  that  the  reset- 
tability errors  of  the  station-selecting  device  and  the  oscillator  drift  can  be  corrected  by 
the  user. 

OSCILLATOR  INJECTION.  The  oscillator  signal  is  injected  into  the  modulator  grid 
circuit  by  either  magnetic  or  capacitive  coupling.  In  many  cases,  the  stray  capacitance 
of  the  station-selector  circuit  wiring  is  sufficient. 

26.  PICTURE  I-F  AMPLIFIER 

The  picture  i-f  amplifier  provides  most  of  the  amplification  required  and  also  controls 
the  frequency  band  of  the  signal  in  the  picture  channel.  The  pass  band  varies  in  width 
from  about  2  Me  in  a  low-definition  receiver  to  about  4  Me  in  a  high-definition  receiver. 
The  i-f  amplifier  attenuates  signals  on  the  adjacent  channels  so  that  thes3  signals  do  not 
interfere  with  the  picture. 

FREQUENCY.  The  choice  of  the  frequency  band  for  i-f  amplification  is  governed 
largely  by  interference  from  direct  i-f  pick-up  and  image  signals.  The  local  oscillator  is 
usually  located  on  the  high-frequency  side  of  the  signal  as  this  simplifies  the  image  inter- 


20-50 


TELEVISION 


ference  problem,  and  thus  the  picture  intermediate  frequency  is  higher  than  the  sound 
intermediate  frequency.  A  choice  of  at  least  20  Me  for  the  sound  intermediate  frequency 
eliminates  other  television  stations  as  images,  but  the  f-m  broadcast  stations  are  then  in 
the  image-signal  range.  A  choice  of  29  Me  or  higher  eliminates  both  television  and  f-m 
broadcast  stations  as  image  signals. 

The  approximate  frequencies  of  the  i-f  band  are  chosen  with  regard  to  image  signals, 
and  the  exact  frequencies  are  chosen  to  eliminate  serious  interference  by  direct  i-f  pick- 
up. Of  the  sources  of  strong  signals  in  the  bands  cited  above,  amateur  frequencies  are 
worst,  since  the  transmitter  may  be  close  to  the  receiver. 

Most  current  designs  utilize  a  frequency  of  about  26  Me  for  the  picture  i-f  carrier. 

VESTIGIAL  SIDEBAND  REQUIREMENTS.  For  band-width  conservation,  picture 
signals  are  transmitted  by  a  vestigial  sideband  system  (reference  66.  See  also  article  9). 
In  this  transmission  system,  it  is  necessary  to  attenuate  the  carrier  frequency  6  db  relative 
to  the  transmission  of  the  major  sideband,  and  to  adjust  the  transmission  of  both  sidebands 
adjacent  to  the  carrier  so  that  a  uniform  output-frequency  spectrum  results  when  a  uni- 
form input  spectrum  is  applied  to  the  system.  This  adjustment  is  made  in  the  receiver. 

In  most  receivers,  the  carrier  and  low  video-frequency  transmission  is  equalized  prior 
to  detection  as  shown  in  terms  of  the  r-f  signal  by  B  of  Fig.  20,  Article  9.  Over  a  frequency 
band  of  about  1.5  Me,  the  transmission  drops  from  full  value  to  10  per  cent  or  less,  with 
the  cutoff  characteristic  so  chosen  that  the  carrier  is  transmitted  at  50  per  cent  of  full 
transmission. 

It  is  desirable  that  this  cutoff  characteristic  be  as  gradual  as  the  standards  allow.  This 
reduces  the  distortion  produced  by  the  quadrature  component  (references  51,  22,  and  23) 
and  the  non-linear  phase  characteristic  associated  with  the  amplitude  cutoff. 

ATTENUATION  CHARACTERISTIC.  A  typical  picture  i-f  amplifier  transmission 
characteristic  is  shown  in  Fig.  3.  The  attenuation  of  the  desired  sound  carrier,  which  is 


23  24  25          26  27 

Intermediate  frequency,  me 

FIG.  3.     Typical  Receiver  Picture  I-f  Response 


28 


normally  obtained  by  the  use  of  traps  in  the  i-f  coupling  impedances,  should  not  be  too 
steepsided,  since  a  high  cutoff  slope  will  convert  the  sound-frequency  modulation  to  am- 
plitude modulation,  which  may  show  in  the  picture.  The  sound  attenuation  should  be 
about  300  kc  wide,  3  db  above  the  minimum,  so  that  frequency  drift  of  the  local  oscillator 
will  not  cause  sound  interference  in  the  picture. 

It  is  desirable  that  about  40  to  50  db  total  attenuation  be  provided  for  the  sound  car- 
rier. This  may  be  produced  entirely  by  i-f  selectivity,  or  it  may  be  obtained  partly  in  the 
i-f  amplifier  and  partly  by  attenuation  at  4.5  Me  in  the  video  amplifier. 

Present  station-assignment  practice  is  such  that  adjacent  channels  will  not  be  allocated 
in  any  region.  The  overlapping  service  areas  of  any  two  regions  whose  allocations  occupy 
successive  channels  are  small.  Therefore,  it  appears  reasonable  to  provide  attenuations  of 
about  35  db  minimum  relative  to  the  desired  picture  carrier  for  the  adj  acent  channel  carriers. 

It  is  usually  necessary  to  use  traps  to  secure  the  attenuation  at  the  sound  carrier  of  the 
lower-frequency  adjacent  channel. 

COUPLING  NETWORKS.  Coupling  networks  used  hi  the  i-f  amplifier  are  of  several 
forms:  double-tuned  transformers  (reference  67),  filter-type  networks,  and  stagger-tuned 
resonant  circuits  are  commonly  used.  See  Section  7  for  more  information. 


PICTURE  I-F  AMPLIFIER 


20-51 


Various  methods  are  used  to  incorporate  traps  in  the  i-f  amplifiers.  Traps  may  be 
part  of  the  coupling  impedance,  or  they  may  be  used  to  reduce  the  effective  transcon- 
ductance  of  the  amplifier  tube. 

Figure  4  shows  several  circuits  in  which  traps  are  employed  to  reduce  the  transmission 
in  a  desired  frequency  range.  The  transmission  characteristics  are  also  shown.  Figure 
4A  shows  a  single  stage  in  which  a  stagger-tuned  single  resonant  circuit  is  used.  The  grid 
leak  of  the  following  tube  is  chosen  to  provide  the  required  Q.  An  inductively  coupled 


Frequency 


Ji/a 


Frequency 


FIG.  4.     Typical  I-f  Coupling  Networks 


Frequency 


trap  is  used  to  secure  attenuation  at  frequency  /i.  An  undesirable  feature  of  the  in- 
ductively coupled  trap  shown  is  the  spurious  response  at  frequency  fz.  The  magnitude 
of  this  spurious  response  is  proportional  to  the  Q  and  coupling  of  the  trap,  and  its  max- 
imum can  be  substantially  the  same  as  the  main  response. 

Figure  4B  shows  a  single  stage  in  which  the  coupling  impedance  is  a  section  of  an  m- 
derived  filter.  The  response  may  be  made  uniform  over  the  desired  band  with  attenua- 
tion at  a  specified  frequency  /i. 

Figure  4C  shows  a  single  stage  in  which  the  coupling  impedance  is  two  coupled  circuits. 
To  secure  maximum  gain,  the  damping  is  concentrated  on  one  circuit  only.  Attenuation, 
at  frequency  /i  may  be  produced  by  a  parallel  resonant  trap  in  the  second  tube  cathode 
circuit.  Such  a  trap  usually  affects  the  input  impedance  of  the  tube  because  of  feedback 
to  the  grid  circuit.  It  is  undesirable  to  use  cathode  traps  with  tubes  in  which  the  sup- 
pressor grid  is  connected  to  the  cathode.  As  the  cathode  has  considerable  impedance  to 
ground,  the  suppressor-to-anode  capacitance  may  couple  sufficient  signal  from  the  output 
circuit  back  to  the  input  circuit  to  cause  instability. 

As  the  trap  attenuation  is  a  function  of  both  the  transconductance  of  the  tube  and  the 
impedance  of  the  trap,  cathode  traps  are  normally  employed  in  fixed-gain  stages. 


20-52 


TELEVISION 


GAIN  CONTROL.  Gain  control  is  usually  accomplished  by  varying  the  bias  of  one  or 
more  i-f  amplifier  tubes.  It  is  generally  necessary,  because  of  the  use  of  high-trans  con- 
ductance tubes,  and  circuits  in  which  the  tube  capacitance  is  a  large  part  of  the  total  ca- 
pacitance, to  stabilize  the  input  capacitance  of  the  gain-controlled  stages  by  an  unby- 
passed  cathode  resistor  (reference  68) . 


27.  PICTURE  CHANNEL  SECOND  DETECTOR 

The  second  detector  in  the  picture  channel  is  usually  of  the  diode  type.  The  video- 
frequency  output-signal  load  impedance  is  determined  by  the  shunt  capacitance  of  the 
diode  and  the  following  stage  and  by  the  band  width  to  be  transmitted.  If  the  impedance 
band-width  product  is  too  low  when  the  total  capacitance  is  lumped  as  a  single  capacitor 
to  ground,  then  video-filter  technique  may  be  used.  The  total  capacitance  is  then  broken 
up  into  several  smaller  units,  allowing  the  impedance  band-width  product  to  be  increased. 
See  Section  7. 

DIODE  LOAD.  If  the  simple  load  circuit  as  shown  in  Fig.  5A  is  used,  the  rise  time 
for  the  video  output  signal  for  outward  modulation  of  the  carrier  will  generally  be  shorter 


B   * 
FIG.  5.     Picture  Detector  Circuits 

than  the  fall  time  for  inward  modulation  of  the  carrier.  The  diode  and  generator  resist- 
ance shunt  the  diode  load  time  constant  for  outward  modulation  but  not  for  inward  modu- 
lation. 

If  the  diode  load  circuit  comprises  a  filter  network  as  shown  in  Fig.  5J5,  the  rise  and  fall 
times  of  the  video  output  signal  are  more  nearly  equal  because  they  are  determined  by  the 
rate  at  which  energy  propagates  through  the  filter  network.  The  input  should  be  at  the 
open  end  of  the  filter,  since,  with  this  connection,  there  is  minimum  reflection  in  the  filter 
which  would  affect  the  current  supplied  by  the  relatively  low  effective  impedance  of  the 
diode  and  i-f  output  circuit. 

There  may  be  variations  in  the  charging  time  with  the  video  modulating  frequency, 
since  the  output  impedance  of  the  i-f  circuit  driving  the  diode  is  not,  in  general,  constant. 
This  effect  is  usually  not  serious. 

TUBE  CONSIDERATIONS.  The  diode  load  impedance  is  low,  ranging  from  about 
2000  to  8000  ohms.  The  output  voltage  generally  ranges  from  about  1  to  5  volts,  and  this 
results  in  high  peak  currents.  To  minimize  the  signal  loss  in  the  diode,  it  is  thus  desirable 
to  use  a  high-perveance  tube  with  low  interelectrode  capacitance. 

I-F  HARMONIC  INTERFERENCE.  The  use  of  a  four-terminal  diode  load  impedance 
like  that  of  Fig.  5B  generally  attenuates  the  ripple  frequency  component  sufficiently  so 
that  it,  or  its  harmonics,  do  not  produce  spurious  patterns  in  the  picture.  These  i-f 
harmonics  can  be  troublesome  on  channels  where  they  fall  in  the  r-f  picture  frequency 
band,  if  they  are  fed  back  to  the  r-f  section  of  the  receiver  with  sufficient  level,  as  a  beat- 
frequency  signal  lying  within  the  video-frequency  passband  of  the  receiver  will  then  be 
produced.  A  beat  takes  the  form  of  alternate  dark  and  light  bands  in  the  picture.  Since 
the  beat  signal  is  not  related  harmonically  to  the  scanning  rate,  the  bands  continually 
move  about  the  picture.  With  full-wave  rectification  the  fundamental  ripple  frequency 
superposed  on  the  video  signal  is  twice  the  intermediate  frequency  and  it  is  reduced  in 
amplitude,  which  simplifies  the  filtering. 

OUTPUT  POLARITY.  The  diode  detector  may  be  arranged  to  produce  a  video  output 
signal  of  either  polarity.  For  the  negatively  modulated  signals  standardized  in  the  United 
States,  the  detector  circuits  of  Fig.  5  will  deliver  video  output  signals  of  negative  polarity; 
that  is,  the  synchronizing  signals  will  be  the  most  positive  part  of  the  video-frequency 
output. 


VIDEO  AMPLIFIERS  AND   DISPLAY 


20-53 


28.  VIDEO  AMPLIFIERS  AND  DISPLAY 

The  video  amplifier,  in  a  television  receiver,  raises  the  level  of  the  picture-detector  out- 
put signal  to  a  satisfactory  value  for  the  picture  tube.  The  input  level  is  commonly  about 
1  to  5  volts  peak-to-peak,  and  the  output  level  ranges  from  about  20  to  about  100  volts 
peak-to-peak.  It  must  be  remembered  that  the  signal  range  from  black  level  to  the 
synchronizing-signal  peaks  does  not  contain  picture  information.  Thus  the  video  am- 
plifier must  handle  a  complete  signal  about  40  per  cent  larger  than  the  black  to  white 
signal.  One  or  two  stages  of  video-frequency  amplification  generally  suffice  in  the  usual 
television  receiver. 

FREQUENCY  REQUIREMENTS.  The  video  amplifier  transmits  a  wide  band  of  fre- 
quencies, one  cutoff  being  at  some  low  frequency,  which  may  be  direct  current.  The  other 
cutoff  is  at  some  high  frequency,  usually  lying  between  about  2  and  4.5  Me,  depending  on 
the  desired  resolution. 

Direct  coupling  is  difficult  to  use  in  a  multistage  amplifier  because  of  the  problem  of 
obtaining  proper  electrode  potentials.  It  is  simpler  to  design  amplifiers  whose  lower 
cutoff  lies  in  the  range  of  about  10  to  10,000  cycles  per  second.  Effective  transmission  of 
the  d-c  component  of  the  signal  may  be  accomplished  by  means  of  a  locally  generated  d-c 
component.  This  involves  a  d-c  reinserter  as  described  in  article  7. 

•B-f 


.  Two-stage  Video  Amplifier 
-B-h 


.B.  Video  Amplifier  with  D-c  Reinserter 
FIG.  6.     Typical  Video  Amplifiers 

In  Fig.  QA  there  is  shown  a  two-stage  video  amplifier.  D-c  reinsertion  is  provided  by 
the  high  grid-cathode  conductance  for  positive  grid  potentials  of  the  second  amplifier 
tube.  The  picture  tube  is  direct  coupled  to  the  output  of  the  second  video  amplifier  tube. 


20-54  TELEVISION 

A  gain  control  which  operates  by  varying  the  amount  of  negative  regeneration  for  al- 
ternating current  is  provided  in  the  cathode  circuit  of  the  second  stage.  A  change  in  gam 
of  about  6  to  1  may  be  obtained  with  uniform  frequency  response  with  a  control  of  this 
type.  The  rheostat  should  be  non-inductive,  and  its  range  is  limited  by  shunt  capacitance 
•which  by-passes  it  for  high  video  frequencies. 

A  potentiometer  is  sometimes  used  as  an  alternative  gain  control,  in  which  the  signal 
on  the  arm  of  the  control  is  supplied  to  the  amplifier  grid.  In  this  case,  shunt  capacitance 
to  ground  from  the  amplifier  grid  may  vary  the  frequency  response  with  gain-control 
setting. 

Figure  65  shows  a  circuit  in  which  the  signal  is  a-c  coupled  from  the  video  amplifier  to 
the  picture  tube.  A  d-c  reinserter  is  provided  to  stabilize  the  potential  of  synchronizing 
signal  peaks  on  the  picture  tube  grid. 

For  d-c  amplifiers  or  those  using  d-c  reinserters  on  the  input  circuits,  it  is  essential  that 
the  amplifier  stage  have  the  same  gain  for  direct  current  as  for  other  frequencies  within 
its  passband.  This  requires  that  the  cathode,  screen,  and  anode  supply  potentials  be 
stabilized  against  variation  with  varying  direct  current  flowing  in  the  amplifier  tube. 
Failure  to  meet  this  requirement  means  that  the  instantaneous  brightness  of  any  part  of 
the  picture  will  be  dependent  upon  the  average  brightness. 

If  the  band  width  to  be  transmitted  is  not  extreme,  and  the  shunt  capacitance  not  high, 
a  simple  two-terminal  constant-fc  type  of  network  is  often  used  for  a  coupling  impedance. 
This  type  may  be  designed  for  quite  uniform  amplitude  and  phase  characteristics.  When 
higher  impedance  is  desired,  or  the  circuit  is  required  to  work  with  high  total  shunt  ca- 
pacitance, it  is  common  to  use  four-terminal  networks.  This  type  allows  the  shunt  ca- 
pacitance to  be  broken  into  .smaller  lumps,  thus  giving  a  higher  impedance-band  width 
product.  For  maximum  exploitation  of  the  band  width,  the  circuit  is  designed  as  a  filter 
(reference  69).  Such  a  filter,  while  it  provides  a  maximum  of  uniform  amplitude  pass- 
band,  has  a  fairly  sharp  cutoff  characteristic  and  a  non-linear  phase  curve,  both  of  which 
may  produce  objectionable  distortion  in  the  form  of  echoes  (reference  9)  of  the  original 
signal.  In  general,  better  performance  is  obtained  with  a  network  in  which  the  amplitude 
characteristic  falls  gradually  with  increasing  frequency,  as  this  reduces  both  the  phase  and 
amplitude  distortion.  Section  7  gives  more  specific  information  regarding  the  design  of 
coupling  impedances. 

The  video-frequency  coupling  network  is  generally  designed  to  have  uniform  impedance 
if  it  is  of  the  two-terminal  type,  or  uniform  transfer  impedance  if  it  is  of  the  four-terminal 
type.  Such  designs  give  uniform  gain  if  driven  by  high-impedance  sources  but  do  not 
produce  uniform  gain  if  the  driving  source  impedance  approximates  that  of  the  network. 
For  the  two-terminal  type  this  is  true  because  the  network  impedance  is  complex  and  has  a 
variable  phase  angle  over  the  transmitted  band.  For  the  four-terminal  type  this  is  true 
because,  for  uniform  transfer  impedance,  the  input  impedance  at  the  driving  point  is  either 
uniform  in  magnitude  but  complex  with  a  variable  phase  angle  over  the  transmitted  band, 
or  non-uniform  in  both  magnitude  of  impedance  and  phase  over  the  transmitted  band. 

Normally,  video  amplifiers  use  the  grid-cathode  circuit  for  input,  and  the  plate-cathode 
circuit  for  output,  and  thus  the  signal  polarity  is  reversed  in  going  through  a  stage.  In 
the  design  of  a  receiver,  the  picture  detector  must  be  so  poled  that  the  desired  polarity  is 
obtained  at  the  picture  tube  grid. 

PICTURE  TUBE.  The  present-day  picture  tubes  are  of  the  cathode-ray  type.  The 
electron  beam  is  focused  by  an  electron  gun  which  may  utilize  electric  fields  only  or  a  com- 
bination of  electric  and  magnetic  fields.  Deflection  of  the  cathode-ray  beam  is  produced 
by  either  electric  or  magnetic  fields.  In  the  present  state  of  the  art,  magnetic  focus  and 
deflection  appear  to  give  the  best  performance  in  regard  to:  (1)  spot  size;  (2)  high  current 
in  the  beam;  (3)  uniformity  of  focus  over  the  raster. 

For  direct-view  receivers,  the  final  anode  voltages  range  from  about  3  kv  to  about  15  kv. 
For  projection-type  receivers,  the  final  anode  voltage  in  current  designs  is  about  30  kv. 

The  phosphor  produces  a  white  light  which  may  vary  in  shade  from  slightly  bluish  or 
greenish  to  yellowish. 

It  is  usual  to  provide  a  bias  control  to  adjust  the  average  brfghtness  of  the  picture.  Ex- 
amples of  this  are  shown  in  Fig.  6.  Sufficient  bleeder  current  flows  through  the  bias  con- 
trol so  the  picture-tube  current  does  not  vary  the  bias  appreciably,  and  a  by-pass  is  pro- 
vided for  high-frequency  currents. 

Section  15  contains  more  detailed  information  on  picture  tubes. 

PICTURE  GAIN  CONTROL.  Automatic  gain  control  for  the  picture  channel  is  desir- 
able in  television  receivers,  as  it  minimizes  readjustment  of  the  controls  when  switching 
from  one  channel  to  another.  Badiated  signals  conforming  to  the  FCC  standards  include 
the  d-c  component.  Thus,  the  average  carrier  level  is  dependent  upon  the  picture  content 
as  explained  in  articles  7  and  9.  It  is  necessary  that  the  automatic-gain-control  circuit  re- 


NOISE  LIMITERS 


20-55 


spond  to  a  part  of  the  signal  which  is  independent  of  the  transmitted  picture.  With  a  nega- 
tive polarity  signal,  as  prescribed  by  the  FCC  standards,  it  is  most  convenient  to  develop  the 
automatic-gain-control  voltage  from  the  synchronizing  signal  peaks.  This  requires  that 
the  automatic-gain-control  rectifier  load  circuit  have  a  time  constant  of  not  less  than  sev- 
eral lines  duration,  so  that  the  picture  content  cannot  affect  the  automatic-gain-control 
voltage.  A  separate  rectifier  operated  at  the  same  level  as  the  picture  detector  may  be 
used  as  the  source  of  automatic-gain-control  voltage.  Better  performance  may  be  ob- 
tained by  amplifying  the  rectifier  output  voltage  with  a  d-c  amplifier.  Figure  7  shows 


~  -25  v 

FIG.  7.    Picture  Automatic  Gain  Control  Circuit 

such  an  arrangement.  In  this  circuit,  Di  is  the  picture  detector;  DZ  is  a  separate  diode 
with  a  high-impedance  load  having  a  time  constant  of  about  200  jusec.  The  output  of  D2 
is  direct  coupled  to  a  triode  Fi,  the  cathode  of  which  returns  to  a  negative  potential,  in 
this  case,  25  volts.  As  the  signal  level  increases,  the  anode  of  Y\  falls  in  potential,  pro- 
viding an  amplified  voltage  which  is  suitable  for  an  automatic-gain-control  bias.  An  al- 
ternative of  this  circuit  may  be  used,  in  which  the  additional  amplification  occurs  prior 
to  the  automatic-gain-control  rectifier.  This  amplification  may  take  place  at  intermediate 
frequency  or  video  frequency. 


29.  NOISE  LIMITERS 

Noise  Kmiters  are  sometimes  used  to  reduce  the  effects  of  impulse  noise  interference 
upon  the  picture-tube  signal  and  upon  the  synchronizing  performance.  It  is  desirable  to 
limit  the  impulse  noise  to  a  level  no  greater  than  that  of  the  synchronizing  signal  peaks  so 
that  the  operating  bias  of  the  video  amplifier  or  the  synchronizing  signal  separator  is  not 
changed.  In  circuits  where  direct  coupling  is  used,  impulse  noise  generally  does  not 
greatly  affect  the  operating  characteristics.  A-c  coupled  circuits  are  usually  affected  con- 
siderably by  noise. 

Diodes  connected  in  shunt  or  series  in  the  video  amplifier  have  been  used  as  impulse 
noise  limiters  An  example  of  a  shunt-connected  diode  limiter  is  shown  in  Fig.  8.  As  the 


-J-B 


-1C 


To  synchronizing 
signal  separator 


FIG.  8.     Impulse  Noise  Limiter 


20-56  TELEVISION 

video  amplifier  is  direct  coupled  to  the  picture  detector,  its  operating  conditions  are  not 
seriously  affected  by  the  noise.  The  limiter  diode  D\  is  connected  in  its  anode  circuit. 
The  d-c  reinserter  for  the  picture  tube,  and  the  synchronizing  signal  separator,  are  actuated 
by  the  signal  following  the  limiter  and  thus  operate  with  increased  reliability.  The 
limiter  shown  adjusts  itself  to  the  signal  level  and  normally  limits  the  peak  of  each  syn- 
chronizing pulse  slightly.  If  the  noise  has  a  high  duty  cycle,  this  type  of  limiter  will  fail, 
as  the  noise  will  then  begin  to  bias  the  diode  Di  off,  since  resistor  R\  will  not  be  able  to  re- 
move the  added  charge  from  C\  quickly  enough.  By  stabilizing  the  potential  of  the  anode 
of  DI  with  a  bleeder,  the  limiter  will  handle  noise  of  high  duty  cycle  but  will  not  adjust 
itself  to  the  signal  level. 

30.  SOUND  AMPLIFIERS 

I-F  CIRCUITS.  The  sound  i-f  amplifier  of  a  television  receiver  must  provide  adequate 
gain  with  proper  selectivity  characteristics  for  the  f-m  sound  signals  which  accompany  the 
picture.  The  design  features  of  television  receiver  sound  i-f  amplifiers  depend  largely 
on  the  receiver  type,  whether  broadcast  a-m  or  f-m  services  are  to  share  the  channel,  and 
the  amount  of  gain  provided  by  the  circuits  which  precede  the  point  of  sound  i-f  take-off. 

Gain  Requirements.  The  sound  circuits  should  be  capable  of  providing  a  comfortable 
audio  output  with  30  per  cent  modulated  sound  carriers  from  6  to  10  db  weaker  than  the 
threshold  picture  level.  Where  manual  or  automatic  picture-gain-control  circuits  can 
reduce  the  amplification  in  the  overall  sound  channel  by  operating  on  tubes  ahead  of  the 
sound  take-off  point,  an  additional  margin  of  sound  gain  is  required. 

With  present  techniques,  the  threshold  picture  level  is  of  the  order  of  50  juv.  An  over- 
all sound  sensitivity  of  about  10  juv  would  therefore  seem  suitable  for  television,  although 
an  additional  20  db  might  be  desirable  for  broadcast  frequency  modulation. 

When  switched  for  television,  receiver  front-end  circuits  seldom  develop  more  gain  than 
is  lost  by  the  modulator  in  converting  to  the  intermediate  frequencies.  The  sound  i-f 
sensitivity  on  the  modulator  grid  is  therefore  about  the  same  as  the  overall  sound  sen- 
sitivity. The  amount  of  sound-channel  gain  that  may  be  provided  between  the  modu- 
lator grid  and  the  point  of  sound  take-off  varies  widely  with  receiver  designs.  Some  pic- 
ture i-f  amplifiers  can  provide  between  40  and  50  db  of  sound-channel  gain  in  the  modu- 
lator and  first  one  or  two  common  stages.  The  most  serious  objection  to  this  arrangement 
is  the  conflict  that  results  from  manual  or  automatic  picture  gain  control  of  these  stages. 

The  output  of  the  sound  i-f  usually  feeds  either  a  ratio-type  f-m  detector  or  a  limiter 
working  into  a  conventional  f-m  detector.  The  minimum  i-f  output  required  depends  on 
the  detector  and/or  limiter  design  and  is  usually  in  the  range  of  1  to  3  volts. 

Selectance  Characteristics.  The  sound  i-f  amplifier  of  a  television  receiver  must  be 
broad  enough  not  only  to  pass  the  sidebands  of  the  carrier  with  full  25-kc  deviation  but 
must  also  pass  this  signal  when  the  local  oscillator  is  detuned  because  of  drift  or  inaccurate 
resett ability  of  the  tuning  device.  Minimum  6-db  band  widths  between  200  and  400  Kc 
are  usual. 

For  television  service,  the  sound-channel  selectors  should  provide  at  least  20-db  at- 
tenuation at  the  picture  carrier  and  50  db  or  more  against  signals  on  adjacent  channels. 
This  is  considerably  less  severe  than  the  requirements  for  broadcast  frequency  modulation 
as  outlined  in  Section  8.  Where  dual  service  is  contemplated,  the  selectivity  requirements 
should  be  based  on  the  broadcast  freqiiency  modulation,  and  it  is  then  necessary  to  keep 
the  local  oscillator  frequency  drift  within  the  band  width  provided. 

Sound  Take-off  Methods.  Television  receivers  which  pass  3.5-  to  4-Mc  video  band 
width  usually  require  traps  to  provide  sufficient  attenuation  of  the  sound  intermediate 
frequency  in  the  picture  channel.  Such  traps  usually  build  up  a  sound  i-f  voltage  or  cur- 
rent, and  they  may  be  coupled  either  directly  or  through  additional  circuit  elements  into 
the  grid  of  the  first  sound  i-f  amplifier.  This  method  is  applicable  to  simple  coupled  traps, 
to  cathode  traps,  and  to  coil  arrangements  in  stages  coupled  by  filter  circuits. 

Receivers  passing  less  video  band  width  may  not  require  sound  traps.  The  sound  take- 
off may  then  be  from  the  secondary  of  a  transformer  whose  primary  is  connected  in  series 
with  a  picture  i-f  transformer;  or  the  sound  and  picture  i-f  amplifier  grids  may  be  operated 
in  parallel. 

Amplifier  Design.  There  is  usually  negligible  selectivity  for  the  sound  intermediate 
frequency  in  the  common  picture  and  sound  circuits.  Some  selectivity  may  be  designed 
into  the  take-off  circuits.  WTaen  its  gain  and  selectivity  requirements  have  been  estab- 
lished, the  sound  i-f  amplifier  can  be  designed  by  the  techniques  described  in  Section  7. 
An  adequate  number  of  single-  or  double-tuned  circuits  can  give  acceptable  performance 
provided  that  20  to  30  wi  over  stray  capacitance  is  added  to  each  circuit.  The  align- 
ment procedure  will  be  simplified  if  the  circuits  are  under-optimum  coupled. 


SYNCHRONIZATION  20-57 

The  mistiming  of  gain-controlled  stages  should  be  minimized  either  by  means  of  unby- 
passed  cathode  resistors  or  by  tapping  down  the  grid. 

SOUND  DETECTOR  AND  ATJDIO  AMPLIFIER.  The  design  of  the  sound  detector 
and  the  audio  ^  amplifier  for  a  television  receiver  follows  broadcast  f-m  practice  as  dis- 
cussed in  Section  8.  Since  100  per  cent  modulation  on  a  television  sound  carrier  cor- 
responds to  25  instead  of  75  Kc  deviation  as  in  broadcast  frequency  modulation,  only  one- 
third  the  output  voltage  is  obtained  from  equivalent  f-m  detectors.  In  addition,  the  out- 
put performance  of  f-m  detectors  is  usually  degraded  when  the  carrier  frequency  is  in- 
creased, as  in  television  sound  i-f  amplifiers.  The  gain  deficiency  can  sometimes  be  made 
up  by  employing  high-gain  audio  amplifier  circuits,  although  this  is  usually  undesirable, 
as  hum-pickup  difficulties  are  inevitable. 

An  undistorted  electrical  output  of  1  watt  is  probably  adequate  for  many  home  tele- 
vision receivers,  as  the  audience  is  close  to  the  receiver.  Television  receivers  incorporating 
broadcast  f-m  or  a-m  are  usually  capable  of  providing  greater  power  output. 

Since  the  current  drawn  by  an  output  audio  amplifier  varies  with  the  signal,  and  since 
this  current  may  represent  a  sizable  fraction  of  the  total  B  current  drain,  special  considera- 
tion must  be  given  when  video  circuits  obtain  power  from  the  same  B  supply.  Either 
adequate  decoupling  arrangements  must  be  made  or  a  constant  current  output  amplifier 
circuit  must  be  used. 

31.  SYNCHRONIZATION 

Adequate  synchronizing  circuits  are  among  the  most  important  features  that  a  tele- 
vision receiver  must  possess.  The  least  expensive  receiver  must  be  capable  of  synchro- 
nizing on  any  signal  of  reasonable  strength  without  readjustment  of  the  speed  controls. 
More  expensive  receivers  may  be  expected  to  maintain  synchronization  on  threshold  weak 
signals  in  the  presence  of  interference. 

The  procedure  for  effecting  synchronization  in  the  television  receiver  consists,  first,  of 
extracting  the  synchronizing  pulses  from  the  complete  video  wave.  The  line  and  field 
pulses  are  then  usually  separated  from  each  other  and  the  resulting  signals  are  used  to 
synchronize  the  respective  scanning  oscillators.  See  article  10  for  a  discussion  of  this. 

SEPARATION  OF  SYNCHRONIZING  PFLSES.  One  method  for  extracting  the 
synchronizing  information  is  to  provide  a  separate  diode  detector  for  this  purpose.  The 
diode  load  resistor  is  by-passed  by  a  capacitor  proportioned  so  that  the  d-c  voltage  de- 
veloped across  the  load  resistor  cannot  drop  more  than  about  20  per  cent  during  a  line 
interval.  The  charging  current  in  the  capacitor  is  then  a  measure  of  signals  in  excess  of 
80  per  cent  of  the  peak  amplitude  of  the  carrier.  A  voltage  proportional  to  the  charging 
current  may  be  obtained  across  a  small  resistor  in  series  with  the  capacitor. 

This  type  of  separator  exaggerates  amplitude  modulation  of  the  synchronizing  signal 
pulses  which  may  be  present  in  the  complete  signal.  Additional  amplification  and  limiting 
are  usually  required. 

As  full  video  band  width  is  not  required  for  the  synchronizing  pulses,  the  separate  de- 
tector can  be  preceded  by  a  high-gain  narrow-band-width  stage.  If  sufficient  signal  is 
developed,  voltage  across  the  entire  diode  load,  which  is  proportional  to  peak  carrier  am- 
plitude, may  be  used  for  automatic  gain  control  as  described  in  article  28. 

Video  Separation.  Synchronizing  signals  can  be  separated  from  a  composite  picture 
signal  by  the  use  of  limiter  circuits  operated  in  conjunction  with  suitable  d-c  stabilization 
of  the  wave  applied  to  the  limiter.  The  most  frequently  used  limiter  of  this  type  employs 
a  sharp  cutoff  tube,  usually  a  pentode,  with  the  signal  applied  to  the  grid  with  black  pos- 
itive (reference  70).  The  tube  is  usually  operated  with  a  grid  leak  and  blocking  con- 
denser input  circuit,  and  without  bias.  Grid  current  is,  therefore,  drawn  on  the  tips  of  the 
synchronizing  pulses.  The  cutoff  characteristic  of  the  tube  and  the  amplitude  of  the 
applied  video  wave  are  correlated  so  that  the  grid  swing  due  to  the  synchronizing  pulses 
alone  exceeds  the  cutoff. 

The  self-bias  d-c  restoring  method  described  above  results  in  poor  performance  in  the 
presence  of  impulse-type  noise  unless  preceded  by  a  suitable  noise  limiter.  A  strong  noise 
pulse  reaching  the  grid  draws  current  and  depresses  the  grid  wave  until  the  blocking  ca- 
pacitor can  discharge.  Several  synchronizing  pulses  can  thus  be  lost. 

The  synchronizing  information  may  be  extracted  from  picture  signals  of  either  polarity 
by  diode  circuits,  examples  of  which  are  shown  in  Fig.  9.  The  time  constant  of  Ri  and  C 
is  proportioned  so  that  the  voltage  across  R\  drops  to  about  80  per  cent  during  a  line 
interval.  The  diode  then  conducts  only  during  the  synchronizing  pulse.  A  voltage  cor- 
responding to  the  diode's  conduction  is  obtained  across  the  proportionately  smaller  re- 
sistor R2  in  series  with  the  diode. 

The  amplitude  of  the  output  synchronizing  wave  from  diode  separators  of  this  type 


20-58 


TELEVISION 


varies  with  picture  content  and  usually  requires  amplifying  and  limiting  for  good  syn- 
chronization. The  interelectrode  capacitance  of  the  diode  may  couple  high  video-fre- 
quency component  current  into  resistor  #2. 


^-Output 


•^-Output 


Output 


_TL 


FIG.  9.     Diode  Synchronizing  Separator 

SEPARATION  OF  LINE  SYNCHRONIZING  PULSES.  The  line  synchronizing  pulses 
are  usually  separated  from  the  complete  synchronizing  signal  by  differentiation.  A 
typical  differentiating  circuit  is  shown  in  Fig.  13,  p.  20-13.  This  operation  produces  a  wave 
containing  a  series  of  narrow  pulses  coincident  with  the  leading  edges  of  the  equalizing 
pulses,  the  line  synchronizing  pulses,  and  the  broad  field  synchronizing  pulses,  as  shown 
in  Fig.  12,  p.  20-12,  to  assure  continuous  operation  of  the  line  scan  oscillator  throughout 
the  field  retrace  interval. 

SEPARATION  OF  FIELD  SYNCHRONIZING  PULSES.  Field  synchronizing  pulses 
can  be  separated  from  the  complete  synchronizing  signal  by  integration.  To  preserve 
reasonable  rise  time  of  the  output  pulses  and  still  eliminate  the  line  pulses,  a  multistage 
integrator  is  sometimes  used.  An  example  is  shown  in  Fig.  13,  p.  20-13. 

OSCILLATOR  SYNCHRONIZATION.  Triggering.  When  the  triggering  technique  is 
used,  the  oscillators  employed  are  usually  types  that  free-run  at  slower  than  the  correct 
speed  and  the  synchronizing  pulses  are  applied  to  initiate  the  retrace.  The  oscillators 
should  be  designed  to  be  insensitive  to  triggering  except  towards  the  end  of  the  trace  so 
that  the  possible  mistiming  is  limited  to  the  interval  between  the  oscillator's  sensitivity 
to  triggering  and  its  self-retrace.  Multivibrators,  blocking  oscillators,  and  thyratron 
oscillators  are  commonly  used.  Where  oscillator  voltage  appears  on  the  triggering  ter- 
minal, buffers  are  usually  required. 

Good  performance  is  obtained  from  triggered  oscillator  circuits  only  when  the  syn- 
chronizing waves  are  clean.  Video  components  and  other  extraneous  signals  should  be 
small.  The  effects  of  random  noise  can  be  minimized  by  restricting  the  passband  into  the 


SCANNING 


20-59 


synchronizing  circuits.  Impulse  noise  should  neither  greatly  exceed  in  amplitude,  nor 
cause  a  loss  of,  the  synchronizing  pulses  after  the  disturbance. 

Phase  Control.  Phase-controlled  scanning  circuits  employ  oscillators  whose  frequency 
can  be  controlled  by  a  d-c  voltage  (reference  17) .  Scanning  oscillators  can  usually  be  so 
controlled  through  the  use  of  a  d-c  amplifier  or  a  control  tube.  The  control  voltage  is 
obtained  by  measuring  the  phase  difference  between  the  synchronizing  signal  and  a  signal 
from  the  scanning  oscillator. 

The  advantage  in  phase-controlled  synchronizing  is  that  an  extremely  narrow  passband 
can  be  employed  in  the  coupling  between  the  phase  comparison  circuit  and  the  oscillator 
control  point  to  make  the  oscillator  insensitive  to  instantaneous  aberrations  of  the  syn- 
chronizing wave.  Random  noise,  impulse  noise,  and,  usually,  small  amounts  of  video  can 
be  tolerated.  The  use  of  a  sufficiently  narrow  passband  to  achieve  the  desired  degree  of 
stability  tends  to  result  in  a  sluggish  pull-in  characteristic.  Typical  performance  is  to  re- 
quire a  second  or  more  to  lock. 

32.  SCANNING 

Conventional  scanning  circuits  for  television  receivers  usually  employ  scanning  oscil- 
lators and  output  amplifiers.  The  oscillator  output  pulses  are  shaped  as  required  and  ap- 
plied to  the  grids  of  output  amplifiers  (reference  71)  to  produce  voltage  or  current  waves 
of  proper  magnitude  and  shape. 

To  achieve  economies,  the  functions  of  the  oscillator,  wave  shaper,  and  output  amplifier 
are  sometimes  integrated,  as  in  the  circuits  shown  in  Figs.  9  and  11  of  pp.  20-9  and  20-11. 

SAWTOOTH  WAVE-SHAPING  CIRCUITS.  The  voltage  wave  required  on  the  grid 
of  scanning  output  amplifiers  departs  from  being  of  sawtooth  form  only  by  what  is  re- 
quired to  correct  for  the  deficiencies  in  the  output  circuit.  The  usual  method  in  scanning 
generators  is  to  integrate  current  pulses  in  a  capacitor. 

A  typical  sawtooth  generator  is  shown  in  Fig.  10.  The  tube  is  normally  cutoff.  When 
a  positive  pulse  is  applied  to  its  grid,  the  capacitor  in  its  plate  circuit  is  discharged.  Fol- 

B|4- 


Input 


-    C 


•  Output 


4=c 


Input  wave 


Output  wave 


FIG.  10.    Sawtooth  Wave  Generator 

lowing  the  pulse,  the  capacitor  recharges  to  the  supply  voltage  as  shown  at  A.  By  using 
a  time  constant  5  or  10  times  as  long  as  the  interval  between  pulses,  a  reasonably  linear 
sawtooth  wave  may  be  obtained. 


20-60 


TELEVISION 


SCANNING  OSCILLATORS.  Blocking  Oscillators  (reference  72).  The  blocking  os- 
cillator has  been  the  preferred  scanning  oscillator  in  television  receivers.  A  blocking  os- 
cillator is  shown  in  Fig.  11.  When  the  transformer  windings  are  connected  so  that  the 
grid  goes  positive  when  the  plate  goes  negative,  this  circuit  will  start  oscillation  and  will 
generate  a  pulse.  During  the  pulse,  grid  current  flows  and  charges  the  capacitor  C  neg- 
atively, eventually  terminating  the  pulse.  The  capacitor  then  discharges  through  the 
resistor  R  to  initiate  a  new  cycle.  The  free-running  speed  is  controlled  by  the  capacitor- 
the  resistor,  and  the  voltage  BI. 


Plate 

voltage 


Synchronizing 


FIG.  11.     Blocking  Oscillator 

The  blocking  oscillator  may  be  synchronized  by  applying  pulses  to  initiate  conduction 
in  advance  of  the  capacitor  discharge.  For  phase-controlled  scanning  circuits,  the  speed 
of  the  blocking  oscillator  can  be  regulated  by  controlling  the  voltage  BI. 

Multivibrators.  Multivibrators  are  frequently  used  in  low-priced  receivers  for  econ- 
omy. These  circuits  are  generally  regarded  as  being  less  stable,  unless  considerably  more 
than  the  minimum  number  of  essential  parts  are  employed. 

A  multivibrator  circuit  arrangement  useful  for  television  receivers  is  shown  in  Fig.  12. 
The  operation  of  this  multivibrator  is  shown  by  the  wave  forms. 

Since  the  current  in  tube  B  consists  of  a  recurrent  pulse  wave,  a  wave-shaping  circuit 
as  described  in  Fig.  10  may  be  placed  in  the  plate  circuit  of  this  tube  for  generating  a  saw- 
tooth voltage  wave,  as  shown  in  Fig.  12. 

Thyratron  Oscillators.  Thyratron  tubes  filled  with  the  lighter  inert  gases  can  be  used 
as  television  scanning  oscillators.  The  tubes  are  connected  to  discharge  a  capacitor  in  the 
plate  circuit.  Speed  is  controlled  by  varying  either  the  recharge  time  constant  or  by  the 
cathode  bias.  Synchronizing  pulses  can  be  applied  to  the  grid. 

The  advantage  of  the  thyratron  tube  lies  in  its  ability  to  pass  peak  currents  of  high 


SCANNING 


20-61 


amplitude,  but  time  delay  circuits  may  be  required  to  prevent  application  of  anode  voltage 
before  the  cathode  has  reached  proper  operating  temperature. 

OUTPUT  AMPLIFIERS  FOR  ELECTROSTATIC  DEFLECTION.  The  resolving 
capabilities  of  most  electrostatic  receiver  tube  types  can  be  realized  only  when  the  average 
of  the  voltages  on  the  deflecting  plates  of  a  pair  is  maintained  equal  to  the  second  anode 
voltage.  This  necessitates  balanced  deflection  as  well  as  balanced  centering  circuits. 
The  scanning  output  circuits  for  electrostatic  tubes  usually,  therefore,  produce  sawtooth 
waves  of  both  polarities. 

Typical  output  amplifier  circuits  employ  two  voltage-amplifier  tubes  connected  to  give 
opposite  polarity  outputs.  A  separate  phase  inverter  is  seldom  used;  the  second  tube  is 


Output 


Plate 

voltage 

tubeB 


Plate 
voltage 
tube  A 


Cathode 


sthode    fcv 

•ltaee\  h 


h 


Synchronizing 

signal  ,^" 

'"    \ 

Grid  voltage  J 
tube  B 


-Gnd. 


•"Cut  off 
potential 
tube  B 


FIG.  12.     Multivibrator  Scanning  Oscillator  and  Wave  Shaper 

usually  driven  by  the  first,  either  by  the  plate  connection  shown  in  Fig.  8,  p.  20-9,  or 
through  the  common  cathode  resistor.  Where  the  total  scanning  voltage  required  ex- 
ceeds the  order  of  600  volts,  the  B  voltage  required  for  the  output  amplifier  tends  to  be- 
come quite  high. 

OUTPUT  AMPLIFIERS,  MAGNETIC.  Since  most  magnetic  scanning  circuits  operate 
by  generating  and  dissipating  energy  during  each  scan,  the  total  power  and  the  circuits 
for  line  scanning  are  considerably  different  from  those  of  field. 

Linear  magnetic  deflection  is  accomplished  by  passing  a  sawtooth  current  through  the 
windings  of  the  deflection  yoke.  The  internal  resistance  of  the  amplifier,  non-linearity  of 
the  amplifier,  and  impedances  in  shunt  or  in  series  with  the  yoke  usually  require  a  wave 
form  other  than  sawtooth  at  the  amplifier  grid.  The  grid  wave  is  thus  sometimes  ex- 
ponential and  has  a  pulse  component  added  by  inserting  a  resistor  in  series  with  the  ca- 
pacitor in  Fig.  10. 

Line  (reference  82) ,  The  usual  receiver  line  scan  circuit  employs  the  idealized  scanning 
cycle  shown  in  Fig.  10,  p.  20-10.  Where  high  efficiency  is  required,  the  triode  dissipating 
circuit  shown  in  Fig.  11,  p.  20-11,  is  used  either  with  or  without  the  "bootstrap"  connection 
which  reclaims  some  of  the  scanning  energy. 


20-62  TELEVISION 

Diode  circuits  are  shown  in  Fig.  13.  A  high-perveance  diode  may  be  connected  across 
the  yoke  as  shown  in  A.  Lower-perveance  diodes,  having  adequate  voltage  rating,  may 
be  connected  across  the  primary  as  shown  in  B. 


Yoke 


Yoke 


FIG.  13. 


B  + 

Diode  Damping  Circuits 


Field.  The  output  load  on  the  field  amplifier  is  essentially  resistive  so  that  the  con- 
trolled dissipation  circuits  used  in  line  scanning  are  seldom  employed.  Occasionally, 
damping  elements  are  placed  across  the  yoke  to  remove  transients  after  the  retrace.  A 
typical  field  scanning  circuit  is  shown  in  Fig.  14. 


FIG.  14.     Field  Scanning  Circuit 


33.  POWER  SUPPLY 

The  successful  performance  of  a  receiver  is  largely  dependent  on  its  power-supply  char- 
acteristics. When  costs  are  important,  the  problems  of  power  supply  are  among  the  most 
difficult  which  the  receiver  designer  must  face. 

HEATERS.  Coupling.  Undesired  coupling  in  the  r-f  or  i-f  amplifiers  through  the 
heater  wiring  can  be  minimized  by  grounding  one  side  of  the  heaters.  It  is  usually  neces- 
sary to  provide  appropriate  by-passing  of  the  heater  lead  and  to  interpose  occasional  r-f 
chokes. 

Heater-cathode  Potentials.  Except  for  a  few  types,  most  receiving  tubes  have  maxi- 
mum heater-to-cathode  potential  ratings  in  the  neighborhood  of  100  volts.  Since  some 


POWER  SUPPLY  20-63 

television  circuits  employ  tubes  with  cathodes  at  potentials  relative  to  ground  in  excess  of 
this  rating,  it  is  necessary  to  provide  additional  windings  on  the  power  transformer  when 
these  circuits  are  used.  Where  the  off-ground  cathode  is  at  a-c  ground  potential,  it  is 
customary  to  connect  one  side  of  the  heater  winding  or  the  center  tap  to  the  cathode.  If 
the  cathode  is  not  at  a-c  ground  potential,  the  separate  winding  may  be  connected  either 
to  the  cathode  through  a  high  resistor  or  to  a  bleeder  whose  voltage  approximates  the 
operating  potential  of  the  cathode. 

LOW-VOLTAGE  B  SUPPLY.  The  two  major  problems  associated  with  the  low-voltage 
B  supply  for  a  television  receiver  are  power-frequency  ripple  and  cross  coupling. 

Power-frequency  Ripple.  The  entrance  of  the  power-line  frequency  disturbance  in  a 
television  picture  may  cause  horizontal  bands  of  light  and  dark  areas,  narrowing,  widening, 
and  lateral  or  vertical  displacement  of  parts  of  the  picture,  and  defocusing.  These  effects 
may  be  caused  by  the  deflection  of  the  cathode-ray-tube  beam  by  the  magnetic  field  of  the 
power  transformer,  or  by  insufficient  power-frequency  filtering  in  the  power  supply. 

When  a  receiver  is  operated  from  a  power  line  which  is  non-synchronous  with  the  field 
scanning,  the  "hum"  pattern  drifts  upwards  or  downwards  and  is  very  objectionable.  It 
appears  essential  that  commercial  receivers  be  so  designed  that  reception  under  these  con- 
ditions remains  unimpaired,  since  programs  are  even  now  being  relayed  over  several  hun- 
dred miles. 

As  the  frequency  of  the  power  line  at  the  receiver  departs  from  synchronism  with  the 
field  scanning,  the  slowly  drifting  "hum"  pattern  jitters  and  flickers.  This  is  most  ob- 
jectionable in  even  the  slightest  amounts.  For  this  reason,  "hum"  disturbance  in  receivers 
intended  for  operation  from  power  sources  whose  frequencies  are  not  synchronous  with  the 
field  scanning  must  not  exceed  about  2  per  cent. 

The  technique  for  securing  adequate  hum  filtering  is  to  provide  inductance-capacitance 
filters  in  high-current  supplies  and  resistance-capacitance  niters  in  low-current  supplies. 

Magnetic  coupling  into  the  picture  tube  is  most  easily  cured  by  placing  the  power  supply 
at  a  sufficient  distance.  When  this  is  impractical,  shields  of  high-permeability  material 
are  placed  about  the  tube. 

Cross  Coupling.  Unless  precautions  are  taken,  the  B  supply  of  a  television  receiver 
can  be  a  troublesome  source  of  undesired  coupling  between  the  various  circuits.  Tubes 
taking  high  peak  currents  modulate  the  B  supply,  which  may  disturb  circuits  that  handle 
low-level  signals.  To  minimize  this  trouble,  multiple  power  supplies,  large  filter  output 
capacitors,  such  as  50  to  100  /if,  multisection  filters,  and  separate  filters  for  susceptible 
circuits  are  used. 

HIGH-VOLTAGE  SUPPLIES.  The  television  receiver  high-voltage  supply  provides 
the  power  required  by  the  second  anode,  and,  in  some  cases,  the  focus  electrode,  of  the 
cathode-ray  tube.  The  common  types  rectify  power-line  frequency  voltage,  a  separately 
generated  sine-wave  voltage,  or  the  voltage  surge  present  during  the  retrace  of  the  line- 
scanning  oscillator.  The  voltages  required  range  from  2  Kv  for  small  direct-viewed  tubes 
to  30  Kv  for  projection  tubes.  The  useful  current  drain  from  a  high-voltage  supply  may 
vary  between  a  few  microamperes  with  a  dark  screen  to  about  1  ma  for  a  bright  picture. 
Reasonably  good  voltage  regulation  is  required  over  the  black-to-white  current  range  if 
noticeable  change  in  picture  size  and  defocusing  is  to  be  avoided. 

To  operate  successfully  over  a  number  of  years,  the  high-voltage  power  supply  must  be 
designed  to  withstand  the  voltage  it  generates.  The  spacing  around  parts  at  the  high  po- 
tential must  be  sufficient  to  prevent  sparkover  and  the  formation  of  corona  under  unfavor- 
able atmospheric  conditions.  Insulating  materials  should  be  non-carbonizing  so  that  they 
are  not  damaged  by  a  single  flashover  and  do  not  become  semi-conducting  after  periods  of 
service.  Insulating  paths  should  be  sufficiently  long  to  prevent  excessive  leakage  under 
conditions  of  dust  and  high  humidity.  Where  organic  materials  must  be  used,  as  in  a 
power  transformer  or  paper  capacitor,  long  life  can  be  assured  only  if  these  components 
are  impregnated  with  a  good  dielectric  fluid  and  are  hermetically  sealed. 

Power-line  Frequency  Supplies.  This  type  is  commonly  used  where  the  high  voltage 
required  does  not  exceed  4  or  5  Kv.  A  single  section  TT  filter  comprising  two  capacitors 
and  a  resistor  is  used  to  remove  the  power-frequency  ripple.  The  resistor  is  usually  made 
as  large  as  the  voltage  regulation  requirements  will  permit  and  the  capacitors  as  small  as 
will  still  afford  adequate  filtering.  The  capacitors  required  are  usually  between  about 
0.03  and  0.1  ptf-  Since  considerable  energy  is  stored  in  such  capacitors  when  charged  to 
voltages  in  excess  of  a  kilovolt,  these  supplies  can  be  lethal  and  thus  are  not  suitable  for 
home  use  unless  adequate  safety  precautions  are  taken.  These  include  the  following: 

1.  Compartmenting  the  power  supply. 

2.  Interlocks  to  prevent  access  to  the  power-supply  compartment  when  the  power  is  on. 

3.  Bleeders  capable  of  discharging  the  high-voltage  capacitors  to  a  safe  voltage  within 
a  second  or  less. 


20-64  TELEVISION 

4.  A  substantial  connection  between  the  receiver  chassis  and  ground. 

Locally  Generated  Sine-wave  Supplies  (references  73  and  83).  This  type  is  frequently 
jsed  either  where  no  a-c  power  line  is  available  or  where,  for  the  higher  voltages,  the  cost 
:>f  the  transformer,  capacitors,  and  safety  features  of  a  power-line  frequency  supply  are 
excessive.  It  comprises  a  sine-wave  oscillator  usually  operating  in  the  low  r-f  range,  a 
step-up  winding,  and  one  or  more  vacuum  rectifiers  in  either  half-wave  or  voltage  multi- 
plying circuits.  The  rectifier  filaments  are  customarily  powered  from  the  oscillator  by 
windings  on  the  step-up  transformer.  Adequate  filtering  is  usually  obtained  with  a  TT 
filter  comprising  two  capacitors  and  a  resistor,  but  the  capacitors  are  only  a  few  hundred 
micromicrofarads.  When  the  energy  storage  in  the  capacitors  is  kept  low,  this  type  of 
supply  can  be  safe  even  though  exposed  and  usually  requires  only  sufficient  shielding  to 
prevent  interference  by  the  locally  generated  wave. 

Voltage  Surge  Supplies  (IT.  S.  Patent  2,051,372).  This  type  of  power  supply  rectifies 
the  voltage  surge  across  an  inductor  when  the  magnetic  field  surrounding  the  inductor  is 
suddenly  changed. 

The  voltage  surge  type  of  supply  is  used  in  some  receivers  employing  a  magnetically  de- 
flected picture  tube.  A  high-voltage  winding  on  the  line  scan  output  transformer  yields 
the  required  voltage  surge  during  the  scanning  retrace.  This  is  rectified  and  filtered,  and 
the  resultant  voltage  is  applied  to  the  second  anode  of  the  picture  tube.  It  is  necessary  to 
supply  added  scanning  power  when  the  second  anode  power  is  thus  extracted. 

Picture  width,  when  using  a  high-voltage  supply  as  described  above,  is  adjusted  by 
changing  the  current  through  the  yoke  while  not  disturbing  the  currents  in  the  trans- 
former. Otherwise  the  scanning  power  and  the  second  anode  potential  vary  together  and 
no  significant  change  in  picture  width  results. 


OTHER  FORMS  OF  TELEVISION 

By  A.  V.  Loughren 

The  material  of  articles  2-33  relates  primarily  to  monochrome,  monocular  television  of 
sharpness  acceptable  for  home  entertainment,  for  use  with  power-supply  systems  of  60- 
cycle  frequency.  A  change  in  any  of  these  requirements  may  affect  significantly  the  de- 
sign of  the  entire  television  system.  The  more  important  examples  of  such  changed  re- 
quirements include:  (a)  television  standards  of  foreign  countries;  (6)  theater  television; 
(c)  color  television;  (d)  binocular  or  stereoscopic  television;  (e)  television  for  special  uses 
(e.g.,  military,  industrial);  and  (/)  use  of  a  common  transmitter  carrier  for  picture  and 
sound  modulations. 

34,  TELEVISION  STANDARDS  OF  FOREIGN  COUNTRIES 

With  respect  to  power-supply  frequency,  the  practices  of  the  several  countries  dif- 
fer. In  addition,  since  there  has  been  no  attempt  at  international  standardization,  in- 
cidental and  unnecessary  differences  in  standards  exist.  Tbe  standards  used  by  the 
British  Broadcasting  Corporation  (adopted  in  1937  and  reaffirmed  in  1944)  illustrate  both 
points.  They  include: 

(a)  Picture  repetition  rate:  25  cycles  per  second  (standard  power-supply  frequency  is 
50  cycles). 

(6)   Lines  per  frame:  405. 

(c)  Polarity  of  picture  modulation:  positive. 

(d)  Form  of  picture  modulation :  amplitude,  double  side  band. 

(e)  Sound  modulation:  amplitude. 

(/)  Sound  carrier  location:  3.5  Me  below  the  picture  carrier  frequency. 

Comparison  may  be  made  with  the  American  standards,  tabulated  in  article  10.  In 
most  other  respects,  BBC  standards  do  not  differ  significantly  from  those  of  the  United 
States  (references  74  and  75) . 

35.  THEATER  TELEVISION 

Requirements  for  theater  television  differ  from  those  for  home  television  primarily  in 
the  following  respects: 

Highlight  Brightness.  Motion-picture  practice  provides  highlight  brightnesses  of  2 
to  20  ft-lamberts;  the  house  is  dimmed  sufficiently  to  make  this  acceptable. 


COLOR  TELEVISION  20-65 

Resolution.  Pictures  projected  from  35-mm  film  have  resolution  considerably  exceeding 
that  of  television  with  a  4-Mc  band  width.  There  is  some  doubt  that  this  factor  out- 
weighs the  practical  advantage  of  common  standards,  especially  in  view  of  Fig.  1  of 
article  1. 

Operator.    Availability  of  a  trained  operator. 

Picture  Size.     Motion-picture  screens  range  up  to  20  ft  in  width. 

Source  of  Picture.  Provision  will  probably  be  required  for  displaying  pictures  whether 
originally  picked  up  for  broadcasting  or  picked  up  specifically  for  a  chain  of  theaters.  This 
indicates  the  desirability  of  the  common  standards  referred  to  previously. 

36.  COLOR  TELEVISION 

For  reasonably  faithful  reproduction  of  colored  subjects,  each  geometrical  picture  ele- 
ment must  be  represented  not  by  a  single  intensity,  its  brightness  (as  required  for  mono- 
chrome television) ,  but  by  three  separate  quantities.  These  may  be  the  brightnesses  of 
three  "primary"  colors.  Alternatively,  one  signal  may  represent  the  resultant  hue,  a 
second  the  saturation  of  that  hue,  and  the  third,  the  brightness.  Color  measurement 
and  specification  may  be  done  by  either  of  these  alternatives. 

Color  reproduction  systems  are  divided  into  additive  and  subtract ive  systems.  In  an 
additive  system,  for  each  picture  element  an  individual  stream  of  light  energy  in  each  of 
the  primary  colors  reaches  the  eye  of  the  observer.  If  the  light  as  originally  generated  is 
white,  a  major  portion  of  it  is  discarded  in  the  filters,  which  transform  it  to  light  of  the 
primary  colors.  In  a  subtractive  system,  the  originally  produced  white  light  is  modified 
individually  for  each  picture  element  by  subtracting  only  the  unwanted  color  components. 
The  subtractive  system  consequently  shows  an  efficiency  in  the  use  of  a  white-light  source 
which  is  several  times  that  of  an  additive  system.  In  color  photography,  subtractive 
processes  such  as  Technicolor  and  Kodachrome  have  found  much  greater  acceptance  than 
the  additive  processes  such  as  Autochrome  and  Finlay.  In  color  television,  on  the  other 
hand,  only  systems  of  the  additive  type  have  been  developed  sufficiently  to  promise  prac- 
tical utility. 

For  additive  systems,  desirable  reproduction  primaries  are  red,  green,  and  blue,  in- 
dividually chosen  as  compromises  between  purity  (or  saturation  to  increase  the  range  of 
producible  colors)  and  transmission  loss  from  white  light.  Colors  equivalent  to  the  trans- 
missions through  Wratten  filters  numbers  47,  58,  and  25,  when  the  light  source  is  the 
International  Committee  on  Illumination's  Illuminant  "C,"  have  found  some  acceptance 
as  the  red,  green,  and  blue  primaries. 

METHODS  OF  TRANSMISSION.  Numerous  methods  of  transmission  of  color 
television  signals  have  been  proposed.  Of  the  several  ways  of  classifying  these  methods 
a  classification  by  the  time  characteristics  of  the  signal  seems  most  important.  On  this 
basis,  the  systems  may  be  classified  as: 

1.  Simultaneous  systems,  in  which  the  three  elements  of  information  required  to 
describe  a  single  picture  element  are  transmitted  simultaneously. 

2.  Sequential  systems,  in  which  the  three  elements  of  information  are  transmitted 
successively.     The  sequential  systems  which  have  been  proposed  fall  into  the  following 
subclasses: 

(a)  Field  Sequential.  A  complete  picture  field  is  transmitted  in  one  color,  followed  by 
successive  fields  in  the  remaining  colors.  In  a  three-color  system  with  2  :  1  interlace,  six 
fields  must  elapse  before  a  picture  which  is  complete  both  geometrically  and  in  color  may 
be  obtained. 

(&)  Line  Sequential.  A  line  of  one  color  is  transmitted,  followed  by  successive  lines  in 
the  remaining  colors,  and  the  cycle  repeats.  In  this  system  if  the  number  of  colors  is  an 
integral  submultiple  of  the  number  of  lines  in  a  complete  picture,  a  given  picture  line 
(for  example,  the  eleventh)  will  always  be  repeated  in  the  same  color  unless  the  color  se- 
quence switching  is  momentarily  altered  at  the  end  of  a  frame  to  produce  a  new  phasing 
for  the  next  frame.  With  three  colors  it  is  difficult  to  avoid  in  a  line  sequential  system  a 
"crawling"  tendency  in  the  produced  image  which  has  usually  been  exhibited  in  systems 
with  orders  of  interlace  greater  than  2:1, 

(c)  Dot  Sequential.  The  three  bits  of  information  describing  an  individual  picture  ele- 
ment are  transmitted  in  immediate  succession,  after  which  transmission  of  information 
for  the  next  picture  element  takes  place.  In  this  system  a  dot  pattern  somewhat  similar 
to  that  of  a  halftone  engraving  appears  superposed  on  the  colored  portions  of  the  picture. 
It  is  interesting  to  note  that  dot  sequential  systems  are  closer  in  their  characteristics  to 
simultaneous  systems  than  to  the  other  sequential  systems. 

Color  television  systems  may  be  also  classified  in  accordance  with  the  quantities  ex- 


20-66  TELEVISION 

plicitly  represented  by  the  elements  of  the  transmitted  signals.    Among  the  many  possi- 
bilities are  the  following: 

1.  Intensities  of  individual  primary  colors. 

2.  Composite  intensity  (or,  alternatively,  visual  brightness)  plus  two  auxiliary  signals 
representing,  for  example,  the  difference  between  the  apparent  brightness  and  the  red 
and  blue  intensities  respectively. 

3.  Composite  intensity,  hue,  and  saturation. 

PHYSIOLOGICAL  REQUIREMENTS.  Resolution.  It  has  been  shown  that,  in  a 
color  picture,  the  apparent  resolution  is  not  appreciably  impaired  when  the  blue  image  is 
severely  defocused;  it  is  also  known  that  moderate  defocusing  of  the  red  image  is  permis- 
sible. Further,  the  use  of  a  common  signal  to  represent  the  fine  detail  for  all  three 
colors  can  be  demonstrated  as  causing  negligible  impairment  of  picture  quality,  when 
practiced  in  reasonable  amounts. 

Since  brightness  is  largely  determined  by  the  green  content  of  a  color,  it  appears  from 
the  foregoing  facts  that  resolution  may  be  effectively  preserved,  and  yet  the  frequency 
band  effectively  conserved,  by  a  system  of  transmission  in  which  brightness  is  transmitted 
with  a  band  of  several  megacycles,  while  hue  and  saturation  are  transmitted  with  relatively 
narrow  bands.  If  this  practice  is  applied  to  a  system  in  which  separate  components  of 
the  radiated  signal  represent  the  intensities  of  the  primary  colors,  the  low-frequency  por- 
tion of  each  color  component  of  the  signal  will  be  derived  individually  (to  represent  the 
distribution  of  light  of  its  corresponding  color  in  the  subject)  while  the  high-frequency 
portion  will  be  identical  in  all  three  colors.  This  common  high-frequency  signal  has  been 
called  "mixed  highs." 

No  studies  comparable  to  that  of  reference  4  have  been  made  for  color  pictures;  in  the 
absence  of  such  information,  it  seems  probable  that  resolution  comparable  to  that  of  mono- 
chrome television  is  desirable,  especially  for  the  brightness  component. 

Flicker.  The  apparent  brightness  of  a  color  image  is  determined  largely  by  the  green 
component.  If  the  entire  image  (or  an  interlaced  image  .field)  is  produced  in  each  color, 
the  flicker  performance  is  essentially  that  given  by  considering  the  repetition  rate  of  the 
green  component  only. 

Color  Range.  The  available  color  reproduction  range  of  a  television  system  using  the 
reproducing  primaries  suggested  above  is  comparable  to  the  best  ranges  obtained  by 
commercial  color  reproduction  processes  of  other  sorts  such  as  color  film  and  multicolor 
printing.  Such  a  range  appears  adequate. 

COMPATIBILITY.  When  a  color  television  system  is  put  into  use  in  an  area  already 
provided  with  a  monochrome  television  service,  the  question  of  compatability  arises.  A 
color  television  system  is  compatible  with  a  particular  monochrome  television  system 
when  color  television  signals  radiated  by  the  color  system  may  be  received  as  monochrome 
images  of  acceptable  quality  on  the  receivers  of  the  monochrome  system  without  modifi- 
cation to  such  receivers.  The  possession  of  this  characteristic  by  a  color  television  system 
contributes  importantly  to  making  the  commercial  introduction  of  the  color  system  easy 
since: 

1.  Color  transmissions  may  be  started  by  individual  stations  of  the  monochrome  tele- 
vision service  as  soon  as  the  stations  are  equipped  for  color  pickup,  with  no  loss  of  audience 
or  impairment  of  audience  satisfaction. 

2.  Those  viewers  who  are  immediately  interested  in  receiving  color  may  purchase  new 
color  receivers  with  assurance  that,  since  the  audience  for  color  broadcasts  includes  both 
themselves  and  the  monochrome  audience,  program  service  will  develop  rapidly. 

For  a  color  system  to  be  compatible  with  a  monochrome  system  it  must  employ  essen- 
tially the  same  transmission  standards  as  the  monochrome  system.  Any  changes  in  the 
standards  either  must  be  small  enough  in  amount  to  be  without  effect  on  operation  of  the 
monochrome  receivers  or  must  be  in  the  nature  of  additions  to  the  signal  of  a  sort  which 
will  go  undetected  by  the  monochrome  receivers.  Dot  sequential  systems,  because  of 
their  superior  potentialities  for  band-width  economy,  seem  more  likely  to  be  capable  of 
operation  compatible  with  the  present  FCC  monochrome  standards  with  a  satisfactory 
grade  of  reproduced  picture  than  the  systems  using  slower  color  sequence  rates. 

TRANSMITTER.  Color  television  transmitters  differ  necessarily  from  monochrome 
transmitters  in  requiring  the  use  of  a  color  camera  and  possibly  additional  control  signals 
for  color  synchronizing  information,  etc.  In  other  respects,  however,  they  may  be  essen- 
tially similar  to  a  monochrome  transmitter.  The  detailed  requirements  for  the  camera  and 
the  additional  control  signals  are  dependent  on  the  particular  color  television  system 
considered. 

RECEIVER.  A  receiver  for  color  television  differs  necessarily  from  a  receiver  for  mono- 
chrome television  only  in  the  substitution  of  a  display  capable  of  giving  color  reproduc- 
tion and  the  addition  of  circuits  to  convert  the  output  of  the  receiver's  detector  into 


REFEKENCES  20-67 

signals  appropriate  to  the  particular  display.  The  circuits  needed  are  those  used  to 
synchronize  and  phase  the  effective  color  of  the  display  to  correspond  to  the  color  of  the 
bit  of  picture  information  being  supplied  to  the  display  at  the  corresponding  instant.  If 
the  system  is  field  sequential  so  that  the  color  switching  rate  is  once  per  field,  the  circuits 
may  control  the  synchronizing  and  phasing  of  a  motor  driving  a  color  wheel — a  disk  pro- 
vided with  a  series  of  sectors  in  the  successive  colors.  In  an  all-electronic  display  the  color 
signals  must  perform  a  corresponding  function;  in  the  electronic  arrangements  the  rate  of 
switching  from  one  color  to  the  next  may  be  arranged  to  accommodate  either  dot  sequen- 
tial, line  sequential,  or  field  sequential  systems. 
See  also  references  1,  5,  25,  76,  77,  85-92. 

37.  BINOCULAR  TELEVISION 

In  direct  viewing  of  a  scene,  the  images  formed  in  an  observer's  right  and  left  eyes  dif- 
fer. Depth  perception  is  based,  in  part,  on  this  difference,  as  discussed  in  Section  14. 
Presentation  of  suitably  different  reproduced  images  to  an  observer's  two  eyes  will  in 
many  cases  enhance  the  illusion  of  solidity  in  a  reproduced  picture. 

Binocular^  or  stereoscopic  presentation  to  a  viewer  must  provide  that  each  eye  sees  only 
the  picture  intended  for  it.  Among  the  means  used  for  this  purpose  have  been  (1)  bar- 
riers and  lenses  to  direct  each  eye  correctly;  (2)  alternate  display  of  the  right  and  left  images 
with  spectacles  worn  by  the  viewer  containing  a  synchronized  mechanical  shutter;  (3)  al- 
ternate display  of  a  red  right  image  and  a  green  left  image,  with  viewer  spectacles  having 
no  green  transmission  to  the  right  eye  and  no  red  transmission  to  the  left  eye;  and  (4)  al- 
ternate display  of  a  right  image  with  horizontally  polarized  light  and  a  left  image  with 
vertically  polarized  light,  with  complementary  viewer  spectacles. 

For  the  three-dimensional  illusion  to  be  most  effective,  the  spacing  between  the  posi- 
tions of  the  camera  in  taking  the  images  should  be  equal  to  the  average  ocular  separation 
(unless  the  image  is  to  be  magnified  in  reproduction) .  The  presentation  of  separate  pic- 
tures for  the  two  eyes  requires  transmission  of  twice  the  information  (and,  hence,  twice 
the  band  width)  required  for  monocular  presentation. 

The  need  for  special  viewing  devices  has  prevented  any  wide  interest  in  binocular  tele- 
vision. 

38.  TELEVISION  FOR  SPECIAL  SERVICES 

Proposals  for  industrial  and  for  military  uses  of  television  have  often  presented  require- 
ments radically  different  from  those  of  television  broadcasting  for  home  entertainment. 
Examples  of  these  unusual  requirements  have  included:  (1)  effective  protection  against 
jamming;  (2)  secrecy;  (3)  severe  size  and  weight  limitations;  (4)  unattended  transmitter 
and  camera  operation;  (5)  the  high  importance  accorded  to  reliability;  (6)  the  usual  rel- 
atively unfavorable  operating  conditions  (as  compared  to  household  use)  encountered  in 
many  forms  of  industrial  and  military  apparatus  design.  Requirements  of  this  sort  may 
affect  basically  the  design  of  the  television  system,  in  addition  to  their  obvious  effect  on  the 
detailed  design  of  the  apparatus.  For  examples  of  designs  for  military  purposes,  see  refer- 
ence 78. 

39.  DIPLEXING  OF  PICTURE  AND  SOUND 

Use  of  a  single  carrier  for  both  picture  and  sound  modulations  is  of  interest  because  of 
possible  simplification  of  receivers  and  decrease  of  radio  spectrum  space  resulting  from  it. 
Possible  approaches  include:  (a)  use  of  different  (and  mutually  non-interfering)  forms  of 
modulation  for  the  two  signals — for  example,  amplitude  modulation  for  the  picture,  with 
frequency  modulation  for  the  sound;  (6)  sharing  of  time  between  picture  and  sound  signals. 

Method  (a)  has  not  been  tried  widely,  probably  in  view  of  the  general  use  of  vestigial 
sideband  picture  transmission,  with  its  inherent  introduction  of  picture  frequency  modula- 
tion sidebands  representing  all  but  the  lowest  frequencies  and  the  consequent  likelihood 
of  cross-talk  between  the  two  signals  in  a  receiver. 

Method  (b)  has  been  proposed  on  several  occasions  and  has  had  some  laboratory  and 
field  trials.  It  is  found  that  the  frequency  of  the  intervals  in  which  sound  signals  are 
transmitted  is  between  two  and  three  times  the  highest  sound  modulation  frequency  which 
can  be  successfully  transmitted,  and  that  the  resulting  sound-signal-to-noise  ratio  is  de- 
pendent on  the  fraction  of  the  total  time  used  for  sound  and  on  the  details  of  the  sound 
modulation  process.  The  sound  intervals  must  be  so  located  relative  to  the  picture  sig- 
nals, in  time,  as  to  produce  no  visible  effect;  they  may,  therefore,  be  placed  between  the 


20-68  TELEVISION 

synchronizing  signal  and  the  start  of  the  picture  information  for  each  scanning  line.     See 
reference  79  for  a  more  complete  discussion  and  bibliography. 

REFERENCES 

1.  "Optics,"  Section  14  of  this  handbook. 

2.  Wilson,  J.  C.,  Television  Engineering,  Pitman  (1937). 

3.  Engstrom,  E.  W.,  A  Study  of  Television  Image  Characteristics,  Proc.  I.R.E.,  Vol.  21,  No.  12, 

pp.  1631-1651  (December  1933),  and  Vol.  23,  No.  4,  pp.  295-310  (April  1935). 

4.  Baldwin,  M.  W.,  The  Subjective  Sharpness  of  Television  Images,  Proc.  I.R.E.,  Vol.  28,  No.  10, 

pp.  458-468  (October  1940). 

5.  Hardy,  A.  C.,  Handbook  of  Colorimetry,  Technology  Press  (1936). 

6.  Mertz,  P.,  and  F.  Gray,  A  Theory  of  Scanning,  Bell  Sys.  Tech.  J.,  Vol.  13,  No.  3,  pp.  464-515 

(July  1934). 

7.  Wheeler,  H.  A.,  and  A.  V.  Loughren,  The  Fine  Structure  of  Television  Images,  Proc,  I.R.E.,  Vol. 

26,  No.  5,  pp.  540-575  (May  1938). 

8.  Campbell,  G.  A.,  and  R.  M.  Foster,  Fourier  Integrals  for  Practical  Applications,  Bell  Telephone 

System  Monograph  B584. 

9.  Wheeler,  H.  A.,  The  Interpretation  of  Amplitude  and  Phase  Distortion  in  Terms  of  Paired  Echoes, 

Proc.  I.R.E.,  Vol.  27,  No.  6,  pp.  359-385  (June  1939). 

10.  Robinson,  D.  M.,  The  Supersonic  Light  Control,  Proc.  I.R.E.,  Vol.  27,  No.  8,  pp.  483-486  (August 

1939). 

11.  Sieger,  J.,  Television  Receivers  Using  the  Scophony  Optical  Scanning  System,  Proc,  I.R.E.,  Vol. 

27,  No.  8,  pp.  487-492  (August  1939). 

12.  Wikkenhauser,  G.,  Synchronization  of  Scophony  Television  Receivers,  Proc.  I.R.E.,  Vol.  27, 

No.  8,  pp.  492-496  (August  1939). 

13.  Lee,  H.  W.,  Optical  Design  of  Television  Receiver  Using  Moving  Scanners,  Proc.  I.R.E.,  Vol.  27, 

No.  8,  pp.  496-500  (August  1939). 

14.  Rosenthal,  A.  H.,  A  System  of  Large-screen  Television  Reception  Based  on  Certain  Electron 

Phenomena  in  Crystals,  Proc.  I.R.E.,  Vol.  28,  No.  5,  pp.  203-212  (May  1940). 

15.  Donal,  J.  S.,  Jr.,  Cathode-ray  Control  of  Television  Light  Valves,  Proc.  I.R.E.,  Vol.  31,  No.  5, 

pp.  195-208  (May  1943). 

16.  Donal,  J.  S.,  Jr.,  and  D.  B.  Langmuir,  A  Type  of  Light  Valve  for  Television  Reproduction,  Proc. 

I.R.E.,  Vol.  31,  No.  5,  pp.  208-214  (May  1943). 

17.  Wendt,  K.  R.,  and  G.  L.  Fredendall,  Automatic  Frequency  and  Phase  Control  of  Synchronization 

in  Television  Receivers,  Proc.  I.R.E.,  Vol.  31,  No.  1,  pp.  7-15  (January  1943). 

18.  Poch,  W.  J.,  and  D.  W.  Epstein,  Partial  Suppression  of  One  Side  Band  in  Television  Reception, 

Proc.  I.R.E.,  Vol.  25,  No.  1,  pp.  15-31  (January  1937). 

19.  Hollywood,  J.   M.,   Single-Sideband  Filter  Theory  with  Television  Applications,  Proc.  I.R.E., 

Vol.  27,  No.  7,  pp.  457-472  (July  1939). 

20.  Nergaard,  L.  S.,  A  Theoretical  Analysis  of  Single-Sideband  Operation  of  Television  Transmitters, 

Proc.  I.R.E.,  Vol.  27,  No.  10,  pp.  666-677  (October  1939). 
Detai 


21.  Goldman,  Stanford,  Television  Detail  and  Selective-Sideband  Transmission,  Proc.  I.R.E.,  Vol.  27, 

No.  11,  pp.  725-732  (November  1939). 

22.  Kallman,  H.  E.,  and  R.  E.  Spencer,  Transient  Response  of  Single-Sideband  Systems,  Proc.  I.R.E., 

Vol.  28,  No.  12,  pp.  557-561  (December  1940). 

23.  Singer,  C.  P.,  A  Mathematical  Appendix  to  Transient  Response  of  Single-Sideband  Systems,  Proc. 

I.R.E.,  Vol.  28,  No.  12,  pp.  561-563  (December  1940). 

24-  Wheeler,  H.  A.,  The  Solution  of  Unsymmetrical-Sideband  Problems  with  the  Aid  of  the  Zero- 
Frequency  Carrier,  Proc.  I.R.E.,  Vol.  29,  No.  8,  pp.  446-458  (August  1941). 

25.  Goldmark,  Dyer,  Piore,  and  Hollywood,  Color  Television  Part  I,  Proc.  I.R.E.,  Vol.  30,  No.  4, 

pp.  162-182  (April  1942) ;  Goldmark,  Piore,  Hollywood,  Chambers,  and  Rives,  Color  Television 
Part  II,  Proc.  I.R.E.,  Vol.  31,  No.  9,  pp.  465-478  (September  1943). 

26.  Zworykin,  V.  K,  The  Iconoscope,  Proc.  I.R.E.,  Vol.  22,  No.  1,  pp.  16-32  (January  1934). 

27.  Bedford,  A.  V.,  and  J.  P.  Smith,  A  Precision  Television  Synchronizing  Signal  Generator,  RCA  Rev., 

Vol.  V,  No.  1,  pp.  51-68  (July  1940). 

28.  Fink,  D.  G.  (editor),  Television  Standards  and  Practice— NT 'SC,  McGraw-Hill. 

29.  Mertz,  P.,  Television— The  Scanning  Process,  Proc.  I.R.E.,  Vol.  29,  No.  10,  pp.  529-537  (October 

1941). 

30.  Rose,  Weimer,  and  Law,  The  Image  Orthicon — A  Sensitive  Television  Pickup  Tube.  Proc.  I.R.E., 

Vol.  34,  No.  7,  pp.  424-432  (July  1946). 

31.  Beers,  G.  L.,  The  Focusing  Viewfinder  Problem  in  Television  Cameras,  Proc.  I.R.E.,  Vol.  31,  No.  3, 

pp.  100-106  (March  1943). 

32.  Nicoll,  F.  H.,  A  New  Chemical  Method  of  Reducing  the  Reflection  of  Glass,  RCA  Rev.,  Vol.  6, 

No.  3,  pp.  287-301  (January  1942). 

33.  De  Vore  and  lams,  Some  Factors  Affecting  the  Choice  of  Lenses  for  Television  Cameras,  Proc. 

I.R.E.,  Vol.  28,  No.  8,  pp.  369-374  (August  1940). 

34.  Larsen  and  Gardner,  Image  Dissector,  Electronics,  Vol.  12,  p.  24  (October  1939). 

35.  Monfort  and  Somers,  Measurement  of  the  Slope  and  Duration  of  Television  Synchronizing  Im- 

pulses, RCA  Rev.,  Vol.  6,  No.  3,  pp.  370-389  (January  1942). 

36.  Maloff,  I.  G.,  Gamma  and  Range  hi  Television,  RCA  Rev.,  Vol.  3,  No.  4,  pp.  409-417  (April  1939). 

37.  Bamford,  H.  S.,  A  New  Television  Film  Projector,  Electronics,  Vol.  11,  p.  25  (July  1938.) 

38.  Engstrom,  Beers,  and  Bedford,  Application  of  Motion  Picture  Film  to  Television,  RCA  Rev., 

Vol.  4,  No.  1,  pp.  48-81  (July  1939). 

39.  Bedford  and  Fredendall,  Transient  Response  of  Multistage  Video  Frequency  Amplifiers,  Proc. 

I.R.E.,  Vol.  27.  No.  4,  pp.  277-284  (April  1939). 

40.  Bedford  and  Fredendall,  Analysis,  Synthesis,  and  Evaluation  of  the  Transient  Response  of  Tele- 

vision Apparatus,  Proc.  I.R.E.,  Vol.  30,  No.  10,  pp.  440-457  (October  1942). 

41.  Loughlm,  B.  D.,  A  Phase  Curve  Tracer  for  Television,  Proc.  I.R.E.,  Vol.  29,  No.  3,  pp.  107-115 

(March  1941). 

42.  Kell,  Bedford,  and  Kozanowski,  A  Portable  High  Frequency  Square  Wave  Oscillograph  for  Tele- 

vision, Proc.  I.R.E.,  Vol.  30,  No.  10,  pp.  458-464  (October  1942). 

43.  Townsend,  C.  L.,  Contemporary  Problems  in  Television  Sound,  Proc.  I.R.E.,  Vol.  31,  No.  1, 

pp.  3-7  (January  1943). 


REFERENCES  20-69 

44.  Shelby  and  See,  Field  Television,  RCA  Rev.,  Vol.  7,  No.  1,  pp.  77-93  (March  1946). 

45.  fetrieby  and  Wentz,  Television  Transmission  over  Wire  Lines,  Bell  Sys.  Tech.  J.,  Vol.  20,  No.  1, 

pp.  62-81  (January  1941). 

46.  Strieby  and  Weis,  Television  Transmission,  Proc.  J.E.S.,  Vol.  29,  No.  7,  pp.  371-381  (July  1941). 

47.  HanseU,  C.  W    Radio  Relay  Systems,  Proc.  I.R.E.,  Vol.  33,  No.  3,  pp.  156-168  (March  1945). 

48.  Frus,  H.  1  .,  A  Note  on  a  Simple  Transmission  Formula,  Proc.  I.R.E.,  Vol.  34,  No.  5,  pp.  354-356 

(May  1946). 

49.  Parker,  W.   N.,  A  Unique  Method  of  Modulation  for  High  Fidelity  Television  Transmission, 

Crt    -o-Sf^'J'^'^  Vofe  26'  No'  8'  pp'  946~962  (August  1938). 

50.  PJHA  Nordlohne,  Experimental  Short  Wave  Broadcasting  Station  PCJ,  Phillips  Tech.  Rev.,  Vol.  3, 

No.  1,  pp.  17-27  (January  1938). 

51.  Kell  and  Fredendall,  Selective  Side  Band  Transmission  in  Television,  RCA  Rev.,  Vol.  4,  No.  4, 

pp.  425-440  (April  1940). 

52.  grown  G  H.,  A  Vestigial  Side  Band  Filter,  RCA  Rev.,  Vol.  5,  No.  3,  pp.  301-326  (January  1941). 

53.  Buzalski,  T.  J     A  Method  of  Measuring  the  Degree  of  Modulation  of  a  Television  Signal,  RCA 

Rev.,  Vol.  7,  No.  2,  pp.  265-271  (June  1946). 

54.  Lindenblad,   N.  E.,  Television  Transmitting  Antenna,  RCA  Rev.,  Vol.  3,  No.  4,  pp.  387-408 

(April  1939). 

55.  Carter,  P.  S.,  Simple  Television  Antennas,  RCA  Rev.,  Vol.  4,  No.  2,  pp.  168-185  (October  1939). 

56.  Janes,  R.  B.,  and  W.  A.  Hickok,  Recent  Developments  in  the  Design  Characteristics  of  the  Icono- 

scope, Proc.  I.R.E.,  Vol.  27,  No.  9,  p.  535  (September  1939). 

57.  Johnson,  E.  O.,  Development  of  an  Ultra  Low  Loss  Transmission  Line  for  Television,  RCA  Rev., 

Vol.  VII,  No.  2,  pp.  272-280  (June  1946). 

58.  Herold,  E.  W.,  Local  Oscillator  Radiation  and  Its  Effect  on  Television  Picture  Contrast,  RCA  Rev., 

Vol.  VII,  No.  1,  pp.  32-53  (March  1946). 

59.  Herold,  E.  W.,  Superheterodyne  Converter  System  Considerations  in  Television  Receivers,  RCA 

Rev.,  Vol.  IV,  No.  3,  pp.  324-337  (January  1940). 

60.  Herold,  E.  W.,  The  Operation  of  Frequency  Converters  and  Mixers  for  Superheterodyne  Reception, 

Proc.  I.R.E.,  Vol.  30,  No.  2,  pp.  84-103  (February  1942). 

61.  Jones,  M.  C.,  Grounded-Grid  Radio-Frequency  Voltage  Amplifiers,  Proc.  I.R.E.,  Vol.  32,  No.  7, 

pp.  423-429  (July  1944). 

62.  Seeley,  S.  W.,  Effect  of  the  Receiving  Antenna  on  Television  Reception  Fidelity,  RCA  Rev.,  Vol.  II, 

No.  4,  pp.  433-441  (April  1938). 

63.  Herold,  E.  W.,  An  Analysis  of  the  Signal  to  Noise  Ratio  of  Ultra  High  Frequency  Receivers,  RCA 

Rev.,  Vol.  VI,  No.  3,  pp.  302-331  (January  1942). 

64.  Mount  joy,  G.,  Television  Signal-Frequency  Circuit  Considerations,  RCA  Rev.,  Vol.  IV,  No.  2, 

pp.  204-230  (October  1939). 

65.  Tyson,  B.  F.,  A  Preselector  Circuit  for  Television,  Electronics,  Vol.  13,  pp.  23-25  (November  1940). 

66.  Nyquist,  H.,  Certain  Topics  in  Telegraph  Transmission  Theory,  Trans.  A.I.E.E.,  Vol.  47,  pp.  617- 

644  (April  1928). 

67.  Mountjoy,  G.,  Simplified  Television  IF  Systems,  RCA  Rev.t  Vol.  IV,  No.  3,  pp.  299-309  (January 

1940). 

68.  Freeman,  R.  L.,  The  Use  of  Feedback  to  Compensate  for  Input-Capacitance  Variations  with  Grid 

Bias,  Proc.  I.R.E.,  Vol.  26,  No.  11,  pp.  1360-1366  (November  1938). 

69.  Wheeler,  H.  A.,  Wide-Band  Amplifiers  for  Television,  Proc.  I.R.E.,  Vol.  27,  No.  7,  pp.  429-438 

(July  1939). 

70.  Zworyldn,  V.  K.T  and  G.  A.  Morton,  Television  (a  book),  Chapter  15,  p.  465;  Chapter  17,  p.  524, 

Wiley  (1940). 

71.  Puckle,  0.  S.,  Time  Bases,  Chapter  V,  pp.  64-66,  68-69;  Chapter  VIII,  pp.  115-122,  Wiley  (1943) 

72.  Maloff,  I.  G.,  and  D.  W.  Epstein,  Electron  Optics  in  Television,  Chapter  13,  p.  255,  McGraw-Hill. 

73.  Schade,  0.  H.,  Radio-Frequency  Operated  High-  Voltage  Supplies  for  Cathode-Ray  Tubes,  Proc. 

I.R.E.,  Vol.  31,  No.  4,  pp.  158-163  (April  1943). 

74.  Blumlein,  A.  D.,  The  Marconi-E.M.I.  Television  System:  Part  I.  The  Transmitted  Wave-Form, 

J.  I.E.E.,  Vol.  83,  No.  504,  pp.  758-766  (December  1938). 

75.  Report  of  the  Television  Committee,  1943,  Right  Hon.  Lord  Hankey,  Chairman,  His  Majesty's 

Stationery  Office,  London. 

76.  Judd,  Deane  B.,  Introduction  to  Color,  from  Symposium  on  Color,  American  Society  for  Testing 

Materials,  pp.  1-12. 

77.  Friedman,  Joseph  S.,  History  of  Color  Photography,  American  Photographic  Publishing  Company. 

78.  Marshall  and  Katz,  Television  Equipment  for  Guided  Missiles,  Proc.  I.R.E.,  Vol.  34,  No.  6,  pp. 

375-401  (June  1946). 

79.  Fredendall,  Schlesinger,  and  Schroeder,  Transmission  of  Television  Sound  on  the  Picture  Carrier, 

Proc.  I.R.E.,  Vol.  34,  No.  2,  pp.  49-61  (February  1946). 

80.  Sziklai,  G.  C.,  Current  Oscillator  for  Television,  Electronics,  Vol.  19,  No.  9,  pp.  120-123  (September 

1946)  . 

81.  Zworykin  and  Morton,  Television,  Chapter  15,  pp.  471-473,  Wiley  (1940).  ,_.,„_ 

82.  Friend,  A.  W.,  Television  Deflection  Circuits,  RCA  Rev.,  Vol.  VIII,  No.  1  (March  1947). 

83.  Mautner,  R.  S.,  and  O.  H.  Schade,  Television  High  Voltage  R-F  Supplies,  RCA  Rev.,  Vol.  VIII, 

84.  Wendt,  K.  R.,  The  Television  D-C  Component,  RCA  Rev.,  Vol.  9,  No.  1,  pp.  85-111  (March  1948). 

85.  Evans,  Ralph  M.,  An  Introduction  to  Color,  Wiley  (1948).       ^,,     T,T,T>_TTi,^. 

86.  An   Experimental  Simultaneous   Color-Television  System:   Kell,   R.   D.,   Part  I,   Introduction  ; 

Sziklai,  G.  C.,  Ballard,  R.  C.,  and  Schroeder,  A.  C.,  Part  II,  Pickup  Equipment;  Wendt,  K.  R., 
Fredendall  G.  L.  and  Schroeder,  A.  C.,  Part  III,  Radio  Frequency  and  Reproducing  Equip- 
ment; Proc.  I.R.E.,  Vol.  35,  No.  9,  pp.  861-875  (September  1947)  . 

87.  Battison,  John  H.,  Color  Television  Transmission  Systems,  Tele-Tech,  Vol.  8,  No.  10,  pp.  18-20 


88.  Report°  of  ^CC  Color  TV  Demonstrations  at  Washington,  Tele-Tech,  Vol.  8,  No.  11,  pp.  24-26 


89.  Loomil,6  Franklin,  Highlights  of  FCC  Color-TV  Demonstrations,  Tele-Tech,  Vol.  8,  No.  12,  pp. 

90.  New'DirectfonSVcolor  Television,  Electronics,  Vol.  22,  No.  12,  pp.  66-72  (December  1949). 

91.  Boothroyd,  Wilson,  Dot  Systems  of  Color  Television,  Part  I,  Electronics    Vol.  22,  No.  12,  pp. 

88-92  ^December  1949);  Part  II,  Electronics,  Vol.  23    Np..l,  pp   96-99    January  1950). 
92    RCA  Laboratories,  A  Six-Megacycle  Compatible  High-Definition  Color  Television  System,  RCA 
Rev.,  Vol.  10,  No.  4,  pp.  504-524  (December  1949). 


SECTION  21 
ELECTRONIC  CONTROL  EQUIPMENT 


BY 

B.  J.  DALTON 


FUNDAMENTAL  ELECTRONIC  POWER 
ART.  CIRCUITS  PAQB 

1.  Rectifier  Circuits  and  Applications 03 

2.  Non-controlled  Rectifiers 08 

3.  Controlled  Rectifiers  and  Inverters 09 

4.  Thyratron  and  Ignitron  Contactors,  ...      11 

FUNDAMENTAL  ELECTRONIC  CONTROL 
CIRCUITS 

5.  Stabilized  D-c  Control  Power  Supplies. .     14 

6.  Timing  Circuits 14 

7.  D-c  Amplifiers 16 


ART.  PAGE 

8.  A-c  Amplifiers 16 

9.  Regulating  Circuits 17 

10.  Tliyratron  Grid-control  Circuits 19 

COMPLETE  ELECTRONIC  DEVICES 

11.  Electronic  Relays 20 

12.  Resistance-welder  Controls 23 

13.  D-c  Motor  Control 27 

14.  Side-register  Positioning  Control 30 

15.  Process  Controls 32 

16.  System  Stabilisation 32 


21-01 


ELECTRONIC  CONTROL  EQUIPMENT 

By  B.  J.  Dalton 

FUNDAMENTAL  ELECTRONIC  POWER  CIRCUITS 

The  fundamental  purpose  of  any  electronic  rectifier  is  to  convert  alternating  current 
into  direct  current.  Therefore,  it  can  be  considered  as  a  d-c  power  supply,  the  same  as  a 
battery  or  a  motor-generator  set.  There  are,  however,  two  important  differences  between 
rectifiers  and  generators  or  batteries: 

1.  Batteries  supply  a  smooth  d-c  voltage  output;  generators  have  a  number  of  com- 
mutator segments  so  that  the  instantaneous  voltage  at  the  brushes  is  nearly  constant; 
rectifiers,  on  the  other  hand,  generally  consist  of  a  relatively  small  number  of  phases  and 
rectifying  elements  and  therefore  almost  all  of  them  have  a  considerable  amount  of  ripple 
voltage  in  the  output.    The  ripple  voltage  must  be  considered  in  many  applications,  and 
in  particular  those  involving  less  than  6-phase  rectification,  because  the  ripple  voltage 
may  produce  a  current  ripple  in  some  loads  which  will  cause  excessive  heating  not  only 
in  the  load  but  also  in  the  rectifier  transformer  and  the  rectifying  elements.     A  highly 
inductive  load  is  a  very  desirable  rectifier  load,  because  the  inductance  smooths  out  most 
of  the  current  ripple.    Resistance  loads  and  particularly  counter  emf  or  capacitive  loads 
will  result  in  high  rms  currents  which  cause  additional  heating.    'In  some  counter-emf 
type  loads  it  may  be  desirable  to  include  sufficient  reactance  in  the  rectifier  output  circuit 
to  limit  the  peak  current  to  a  reasonable  value.    This  obviates  the  necessity  for  excessive 
rectifying  element  and  transformer  sizes  and  also  minimizes  the  additional  heating  which 
would  otherwise  be  present  in  the  load. 

2.  Batteries  are  inherently  energy-storage  devices  and  have  a  constant  output  voltage; 
generators  not  only  have  a  small  amount  of  energy  stored  mechanically  in  their  rotor 
but  also  are  usually  driven  by  an  a-c  motor  the  speed  of  which  is  reasonably  independent 
of  a-c  line  voltage;  thus  the  generator  has  a  constant  output  voltage.    Rectifiers,  on  the 
other  hand,  have  no  inherent  energy  storage,  and  the  output  voltage  at  any  instant  is 
directly  proportional  to  the  a-c  input  voltage.    Therefore,  if  a  constant  d-c  output  voltage 
is  desirable,  either  the  a-c  input  voltage  must  be  regulated  or  a  regulating  means  must  be 
provided  for  the  output  voltage. 

The  efficiency  of  electronic  rectifiers  is  determined  by  the  losses  in  the  rectifying  ele- 
ments themselves  together  with  transformer  losses,  cathode  heating  losses,  and  miscel- 
laneous auxiliary  losses.  The  power  factor  of  a  rectifier  is  determined  by  the  type  of 
load  and  number  of  phases  and  for  a  controlled  rectifier  by  the  amount  of  phase  retard  as 
well.  An  inductive  load  will  give  the  highest  power  factor;  a  counter-emf  load  will  result 
in  the  lowest  power  factor.  For  a  given  type  of  load,  the  power  factor  is  inversely  propor- 
tional to  the  amount  of  phase  retard,  the  rectifier  being  essentially  a  constant-kva  load 
on  the  power  line. 

One  or  more  of  the  following  factors  will  govern  the  choice  of  the  type  of  rectifier  for  a 
specific  application: 

1.  The  magnitude  of  the  required  d-c  power. 

2.  The  magnitude  of  the  required  d-c  voltage. 

3.  The  magnitude  of  the  required  d-c  current. 

4.  The  required  degree  of  freedom  from  a-c  voltage  ripple  in  the  d-c  output  voltage  or 
current. 

5.  The  effect  of  the  connected  load  on  the  rectifying  elements. 

6.  "The  number  of  phases  and  the  type  of  connection  available  from  the  a-c  power  supply. 

7.  The  magnitude  of  the  a-c  power  supply  voltage. 

8.  The  necessity  for  adjusting  the  output  voltage. 

9.  The  necessity  for  regulating  the  output  voltage  at  a  specific  value. 

10.  Physical  size. 

11.  Cost. 

Electronic  rectifiers  may  be  classified  in  accordance  with  their  controllability  as  (1)  non- 
controlled,  (2)  grid-controlled,  and  (3)  ignitron  or  pool-type-controlled  rectifiers.  (These 
types  will  be  discussed  in  greater  detail  later.)  Also  electronic  rectifiers  may  be  classified 
in  accordance  with  the  type  of  circuit  connection  used. 

21-02 


RECTIFIER  CIRCUITS  AND  APPLICATIONS  21-03 

The  proposed  AIEE  standards  for  pool-cathode  mercury-arc  power  converters  include 
a  list  of  36  standard  rectifier  circuit  connections.  Table  I,  however,  illustrates  eight  circuit 
connections  which  are  representative  of  a  large  number  of  applications,  and  in  particular 
those  applications  in  the  low-  and  medium-power  field.  These  circuits  and  their  uses  are 
described  below. 

1.  RECTIFIER  CIRCUITS  AND  APPLICATIONS 

Half-wave  rectifiers  are  generally  limited  in  their  application  to  low  power  circuits. 
Typical  half-wave  rectifier  applications  are:  (1)  D-c  power  supplies  for  small  electronic 
amplifiers.  In  this  application  a  filter  circuit  is  used  to  store  energy  during  the  half  cycle 
in  which  the  rectifier  element  is  conducting  so  that  a  reasonably  smooth  output  voltage 
can  be  obtained.  ^(2)  D-c  power  for  operating  d-c  coils  of  magnetic  relays.  (3)  Charging 
circuits  for  capacitors.  (4)  D-c  power  supplies  for  the  armature  power  for  small  d-c 
motors.  (5)  Battery  charging  equipment.  Half-wave  rectifiers  are  generally  used  because 
of  their  simplicity  and  relatively  low  cost.  Furthermore  the  half-wave  rectifier  circuit 
can  be  operated  directly  from  a  single-phase  power  supply,  without  an  anode  transformer, 
if  the  load  is  designed  to  match  the  obtainable  output  voltage.  The  half-wave  circuit, 
however,  is  undesirable  in  some  respects  because  the  energy  delivered  to  the  load  in  an 
entire  cycle  must  be  obtained  during  one-half  cycle.  This  reduces  the  transformer  utiliza- 
tion (if  a  transformer  is  used)  and  also  results  in  high  peak  currents  in  the  rectifying  ele- 
ment. In  addition,  this  type  of  rectifier  is  generally  unsuited  for  application  to  highly 
inductive  (i.e.,  iron-core  inductance)  loads.  The  time  constant  of  an  inductive  load  is 
usually  sufficiently  long  so  that  current  will  not  build  up  during  the  half  cycle  in  which 
the  tube  can  conduct.  The  small  amount  of  energy  that  is  transferred  into  the  load  is 
inverted  during  the  non-conducting  half  cycle.  Usually  the  average  current  is  about  10 
to  20  per  cent  of  that  which  would  be  expected  in  a  pure  resistance.  A  capacitor  or  a 
rectifier  tube  is  sometimes  connected  across  an  inductive  load  to  prevent  the  energy  stored 
in  the  inductance  from  being  transferred  to  the  a-c  supply  during  the  normally  non- 
conducting half  cycle.  Thus  the  current  can  build  up  over  a  period  of  time  and  finally 
reach  a  steady-state  value. 

The  diametric  (full-wave)  rectifier  is  used  in  a  large  number  of  low-power  applications. 
Typical  applications  are:  (1)  d-c  power  supplies  for  other  electronic  equipment;  (2)  d-c 
power  supplies  for  magnetic  clutches,  magnetic  chucks,  and  lifting  magnets;  (3)  battery 
chargers;  (4)  d-c  power  for  supplying  the  fields  of  d-c  motors  and  generators;  (5)  d-c  power 
for  the  armature  circuit  of  d-c  motors.  This  rectifier  is  a  relatively  simple  and  inexpensive 
unit.  In  some  applications,  however,  its  usefulness  is  limited  by  the  amount  of  ripple 
present  in  the  output  voltage.  The  desirability  of  obtaining  larger  amounts  of  power  from 
polyphase  power  supplies  usually  limits  its  use  to  applications  involving  less  than  5  kw 
of  d-c  power. 

The  primary  advantage  of  the  diametric,  double-way  (bridge)  rectifier  is  its  ability  to 
supply  high  voltages.  The  peak  inverse  voltage  across  the  rectifying  element  is  only  half 
of  what  it  would  be  in  a  diametric  (full-wave)  circuit.  Thus  a  diametric,  double-way 
rectifier  can  be  designed  to  deliver  twice  the  d-c  voltage  that  a  diametric  rectifier  can 
deliver,  assuming  the  same  rectifying  elements  in  both  cases.  It  also  results  in  high  trans- 
former utilization.  This  rectifier  circuit  has  another  advantage  in  that  it  can  be  operated 
directly  from  a  single-phase  power  supply  without  an  anode  transformer,  provided  that 
the  output  voltage  obtained  this  way  is  of  a  suitable  value. 

Polyphase  rectifiers  are  generally  used  whenever  large  amounts  of  power  are  required. 
The  selection  of  a  particular  polyphase  rectifier  is  largely  a  matter  of  obtaining  the  desired 
output  voltage  and  current  from  existing  or  standard  rectifying  elements.  For  example, 
if  rectifying  elements  of  5-amp  capacity  each  were  available  and  a  15-amp  d-c  output  were 
required,  a  delta  3-phase  wye  rectifier  circuit  could  be  chosen.  Likewise  if  20  amp  of 
direct  current  were  required  using  the  same  rectifying  elements,  a  Scott  4^phase  cross 
rectifier  circuit  could  be  selected.  A  delta  6-phase  star  rectifier  circuit  could  be  selected 
to  obtain  30  amp  of  direct  current  from  the  same  rectifying  elements.  However,  it  might 
be  that  the  use  of  such  a  circuit  as  the  delta  6-phase  star  would  place  a  severe  duty  on  a 
particular  rectifying  element  either  from  the  standpoint  of  the  rms  current  or  from  the 
standpoint  of  the  peak  current.  For  example,  if  an  excessive  peak  or  rms  rectifier  current 
would  exist  in  a  delta  6-phase  star  circuit,  the  delta  6-phase  double-wye  circuit  could  be 
chosen.  This  would  reduce  the  rms  and  peak  currents  by  a  ratio  of  almost  2  to  1.  It  is 
sometimes  possible  by  careful  coordination  of  power  supply  connections  and  load  voltage 
ratings  to  use  a  delta  6-phase  wye  double-way  rectifier  to  eliminate  the  rectifier  trans- 
former. Double-way  rectifiers,  as  mentioned  before,  will  deliver  higher  d-c  voltages,  for  a 
given  peak  inverse  voltage  across  the  rectifying  elements,  than  single-way  rectifiers. 


21-04 


ELECTRONIC  CONTROL  EQUIPMENT- 


RECTIFIER   CIRCUITS  AND   APPLICATIONS 


21-05 


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21-06 


ELECTBONIC  CONTROL  EQUIPMENT 


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21-08  ELECTRONIC  CONTKOL  EQUIPMENT 

Sometimes  the  requirement  of  low  ripple  currents  governs  the  choice  of  the  rectifier  circuit, 
rather  than  the  current  capacity.  For  very  large  rectifiers  above  1000  kw  (total  installed 
capacity) ,  12  or  more  phase  rectifiers  may  be  used  to  minimize  the  effect  of  rectifier  har- 
monics on  telephone  lines. 

EXPLANATION  OF  TABLE  1.  Table  1  consists  of  four  distinct  sections  appearing 
from  left  to  right  as  follows:  (1)  circuit  nomenclature  and  connection  diagrams;  (2)  output 
voltage  wave  shapes,  required  transformer-secondary  voltage  ratings,  and  output-voltage 
ripple  values;  (3)  rectifying  element  current  wave  shapes  (for  resistance  load),  rectifying- 
element  current  values  (for  resistance  load),  and  rectifying-element  peak  inverse  voltages; 
(4)  rectifier-transformer  kva  ratings,  for  both  resistive  and  inductive  loads.  All  data  are 
tabulated  in  terms  of  the  theoretical  no  load  d-c  output  voltage  (Edo)  and  output  cur- 
rent (Id). 

The  usefulness  of  this  table  can  best  be  illustrated  by  means  of  an  example. 

Problem.  To  specify  (a)  the  transformer-secondary  voltage  rating;  (6)  the  transformer- 
secondary  and  primary  kva  ratings;  (c)  the  average,  rms,  and  peak  current  which  the 
rectifying  elements  will  need  to  carry;  and  (d)  the  peak  inverse  voltage  which  will  exist 
across  the  rectifying  elements,  for  a  diametric  (full-wave)  rectifier  to  deliver  10  amp  of 
direct  current  at  250  volts.  Assume  that  there  is  no  voltage  drop  in  the  rectifying  ele- 
ments; also  assume  a  resistance  load. 

Solution:  (a)  Transformer-secondary  voltage  each  side  of  center  tap  =  1.11  X  25QEdo 

=  277.5  volts 

Total  secondary  volts  =  555.0 
(6)  Transformer  secondary  kva  rating  =  1.75  X  d-c,  kw 

_  1.75  X  10  amp  X  250  volts 

1000 

=  4.37  kva 
Transformer-primary  kva  rating  =  1.235  X  d-c  kw 

_  1.235  X  10  X  250 

1000 
=  3.09  kva 

(c)  Rectifying-element  currents 

Average  =  0.5  X  107<j  =  5  amp 

Rms  =  0.786  X  10/d  =  7.86  amp 
Peak  =  1.57  X  101*  =  15.7  amp 

(d)  Rectifying-element    peak   inverse   voltage  =  3.14  X  250^do  =  785    volts.      (The 
actual  ratings  will  need  to  be  increased  to  compensate  for  the  voltage  drop  in  the  rectifying 
elements,  transformer  reactance,  and  the  like.) 

2.  NON-CONTROLLED  RECTIFIERS 

The  types  of  non-controlled  electronic  rectifying  devices  in  common  use  are:  (1)  high- 
vacuum  tubes;  (2)  hot-cathode  gaseous  rectifier  tubes;  (3)  metallic  rectifiers. 

Non-controlled  rectifiers  are  used  in  applications  where  a  fixed  amount  of  d-c  voltage 
is  required.  Applications  for  non-controlled  rectifiers  include  battery  charging,  motor- 
field  excitation,  generator-field  excitation,  magnetic-chuck  excitation,  lifting-magnet 
excitation,  d-c  control  power  supplies,  etc. 

The  high-vacuum  or  kenotron  rectifier  tubes  are  inherently  low-current  tubes,  because 
the  voltage  drop  and  therefore  the  power  loss  in  the  tube  are  proportional  to  the  current 
flowing.  High-vacuum  rectifier  tubes  may  be  classified  as  low  or  high  voltage.  Low- 
voltage  types  are  used  primarily  for  small  amounts  of  d-c  control  power  for  other  electronic 
equipments.  Often  low-voltage  tubes  are  constructed  with  two  rectifying  devices  in  one 
tube,  thus  making  a  single  tube  suitable  for  a  diametric  (full-wave)  rectifier  circuit.  High- 
voltage  types  are  used  in  such  applications  as  dust  precipitators  and  high-voltage  power 
supplies  for  other  electronic  equipment. 

The  outstanding  characteristic  of  hot-cathode  gaseous-type  rectifier  tubes  is  their 
inherently  low  and  constant  voltage  drop  which  results  in  high  efficiency  in  high-current 
applications.  These  tubes  are  available  in  a  range  of  current  ratings  of  0.1  amp  to  20 
amp  and  from  about  100  volts  to  10,000  volts,  peak  inverse  ratings.  Low-voltage  types, 
such  as  are  used  in  battery  chargers,  are  usually  filled  with  argon.  Xenon-filled,  argon- 
and-mercury-vapor-filled,  and  mercury-vapor-filled  tubes  are  used  in  applications  requir- 


CONTROLLED  RECTIFIERS  AND   INVERTERS          21-09 

ing  750,  2000,  and  10,000  volts,  peak  inverse  ratings,  respectively.  Gaseous-type  rectifiers 
will  not  operate  successfully  in  parallel  without  load-balancing  devices,  because  the  tube 
drop  of  two  paralleled  tubes  may  be  slightly  different  and  the  tube  with  the  lower  drop 
will  carry  all  or  most  of  the  load  current.  A  more  complete  discussion  of  hot-cathode 
gaseous-type  rectifier  tubes  of  both  the  non-controlled  and  the  controlled  type  is  pre- 
sented in  Section  4. 

Copper  oxide  and  selenium-type  metallic  rectifiers  are  used  predominantly  in  the  low- 
voltage  high-current  field.  The  general  characteristics  of  these  rectifiers  are  a  relatively 
low  peak  inverse  voltage  per  rectifying  disk  and  a  relatively  high  current  capacity.  Stacks 
consisting  of  one  or  more  cells  can  be  used  in  series  or  in  parallel  to  increase  their  voltage 
or  current  capacity,  respectively.  The  copper  oxide  rectifier  is  older,  from  the  standpoint 
of  general  usage,  than  the  selenium  rectifier.  The  selenium  rectifier  can  operate  at  a  higher 
temperature  than  the  copper  oxide  rectifier,  and  therefore  for  a  given  rating  it  is  a  some- 
what smaller  unit  than  the  copper  oxide  type.  Selenium  rectifiers  are  usually  operated 
nearer  their  breakdown  voltage  rating  than  copper  oxide  rectifiers;  therefore  the  selenium 
rectifier  is  usually  more  subject  to  damage  on  overvoltage.  Other  materials,  such  as  copper 
sulfide,  exhibit  the  same  rectifying  action  as  the  copper  oxide  and  selenium  materials  but 
are  not  as  commonly  used. 

3.  CONTROLLED  RECTIFIERS  AND  INVERTERS 

Controlled  rectifiers  generally  are  used  whenever  it  is  desired  to  adjust  the  d-c  output 
voltage  level  over  a  reasonably  wide  range,  or  when  it  is  necessary  to  regulate  the  output 
voltage  to  compensate  for  changes  in  the  load  current  or  changes  in  the  input  Line  voltage. 
Typical  applications  of  controlled  rectifiers  are:  (1)  adjustable  d-c  voltage  for  motor-  and 
generator-field  supplies;  (2)  adjustable  d-c  voltage  for  d-c  motor  armatures;  (3)  adjustable 
d-c  voltage  for  the  d-c  windings  of  saturable  reactors  which,  in  turn,  control  motor,  light- 
ing, or  resistance  power  circuits;  (4)  adjustable  d-c  voltage  supply  for  testing  of  various 
d-c  devices;  (5)  d-c  power  for  charging  capacitors  in  energy-storage  resistance  welders  at  a 
given  rate  and  to  a  given  voltage. 

In  small  rectifiers,  it  may  sometimes  be  more  convenient  and  economical  to  adjust  the 
rectifier  d-c  output  either  by  adjusting  the  a-c  voltage  input  or  the  d-c  voltage  output  by 
means  of  a  slide-wire  resistor.  Also  in  large  rectifiers  it  may  be  desirable  to  adjust  the 
output  voltage  by  means  of  induction-voltage  regulators,  adjustable  auto  transformers,  or 
saturable  reactors  in  the  a-c  circuit.  If  these  methods  are  used,  the  rectifying  elements 
can  be  of  the  non-controllable  type. 

Grid-controlled  thyratron  rectifiers  provide  greater  flexibility,  faster  response,  and  less 
bulky  control  equipment  than  rectifiers  controlled  in  the  power  circuits.  Furthermore, 
automatic  control  in  larger  sizes  is  generally  more  economical  with  thyratron  control 
than  with  power  circuit  control. 

Thyratron-type  rectifiers  are  always  built  in  a  single  envelope.  Some  thyratron  tubes 
are  controlled  electromagnetically  by  a  plate  on  the  outside  of  the  tube.  By  far  the  most 
common  practice,  however,  is  to  control  the  tubes  electrostatically  with  a  grid  in  the 
electron  path.  Thyratron  tubes  can  be  obtained  in  ratings  as  low  as  approximately  20  ma 
and  have  been  built  in  ratings  as  high  as  100  amp.  (See  Section  4.)  The  maximum  size 
of  standard  tubes,  however,  is  about  12.5  amp.  (High-vacuum  triodes  are  generally  not 
used  in  controlled  rectifier  circuits.) 

Controlled  rectifier  tubes  may  be  connected  in  any  of  the  circuits  shown  in  Table  1. 
Tubes  and  transformers,  however,  should  be  carefully  selected  so  that  the  ratings  are  not 
exceeded.  As  mentioned  earlier,  the  peak  and  rms  currents  in  a  rectifier  will  be  higher 
on  resistance  and  counter-emf  loads  than  on  inductance  loads.  Not  only  must  this  factor 
be  considered,  but  also  consideration  must  be  given  to  the  higher  peak  and  rms  currents 
that  will  result  from  the  use  of  grid  control  on  loads  such  as  d-c  motor  armature  circuits. 
Another  factor  to  be  considered  is  the  unbalance  in  tube  currents  which  may  exist  in 
polyphase  rectifiers  operating  with  rated  current  and  having  a  large  amount  of  phase 
retard.  Figure  1  shows  the  voltage  and  current  wave  shapes  of  a  diametric  (full-wave) 
rectifier  operating  with  varying  degrees  of  phase  retard  on  an  inductive  load,  a  resistive 
load,  and  a  fixed  counter-emf  load  all  having  different  electrical  characteristics.  Although 
a  diametric  rectifier  is  used  here  for  the  sake  of  clarity,  the  same  fundamental  information 
applies  to  other  rectifier  circuits.  Although  three  different  load  classifications  are  shown, 
many  loads  consist  of  various  combinations  of  resistance,  inductance,  and  counter  emf. 

The  load  current  on  a  very  highly  inductive  load  is  nearly  constant  even  with  a  diametric 
rectifier  circuit.  As  the  firing  angle  is  retarded,  the  energy  from  the  inductance  is  trans- 
ferred back  into  the  line  circuit  during  a  portion  of  the  cycle  when  the  tubes  would  be 


21-10 


ELECTRONIC  CONTROL  EQUIPMENT 


normally  non-conducting.  When  the  firing  angle  has  been  retarded  approximately  90°, 
the  inductance  current  is  theoretically  zero.  Practically,  however,  the  90°  firing  point 
will  result  in  a  current  of  approximately  10  to  20  per  cent  of  the  maximum  which  would  be 
expected  as  calculated  by  Ohm's  law. 

In  a  sense  a  purely  resistance  load  is  somewhat  academic  because  this  type  of  load 
seldom  occurs  in  practice.  It  is,  however,  a  logical  stepping  stone  in  estimating  what  is 
to  be  expected  from  counter-emf  loads.  It  can  be  observed  from  Fig,  1  that  the  load 
current  hi  a  resistance  load  follows  identically  the  wave  shape  of  the  output  voltage. 

Counter-emf  loads  fall  into  two  general  classifications:  fixed  and  variable.  A  battery 
is  a  typical  example  of  a  fixed  counter-emf  load.  The  peak  value  of  load  current  in  a  fixed 
counter-emf  load  is  determined  by  the  difference  between  the  peak  a-c  voltage  and  the 
counter-emf  potential,  and  the  circuit  resistance.  Exceedingly  high  current  can  be  ob- 
tained on  low-impedance  circuits  with  a  very  small  difference  in  voltage  between  the  load 


Firing 
full  on 


inductive  load 


Resistance  load 


Typical  fixed  counter-emf  load; 


FIG.  1.     Rectifier  Output  Voltage  and  Load-current  Wave  Forms  for  Different  Types  of  Loads  and 
with  Several  Degrees  of  Phase  Retard  with.  Diametric  (Full- wave)  Rectifier 

and  the  peak  rectifier  output.  The  inherent  circuit  impedance  may  reduce  the  peak  cur- 
rent to  a  reasonable  value;  if  not,  external  resistance  or  inductance  can  be  added  for  that 
purpose.  Figure  1  shows  that  as  the  firing  angle  of  a  rectifier  on  a  fixed  counter-emf  load 
is  retarded  there  is,  up  to  a  certain  point,  no  change  hi  the  output  current,  but,  beyond 
the  point  where  the  transformer-secondary  voltage  and  the  counter-emf  voltage  intersect, 
a  reduction  in  current  is  obtained  by  a  further  retard  in  firing  position.  A  d-c  motor  arma- 
ture is  a  typical  variable  counter-emf  load.  This  type  of  load  is  quite  different  in  its  opera- 
tion from  a  rectifier  or  other  fixed  counter-emf  load.  A  motor  armature  is  a  good  example 
of  a  load  which  includes  resistance,  inductance,  and  variable  counter  emf . 

The  thyratron  rectifier  circuits  which  have  been  described  can  also  be  used  for  inverters 
to  convert  d-c  power  into  a-c  power,  provided  that  both  d-c  and  a-c  power  supplies  are 
available  for  the  transfer  of  power  and  that  the  grid-control  circuits  are  properly  arranged. 
Many  motor-control  rectifiers  are  operated  as  inverters  during  reversal  of  motor  armatures 
and  during  fast  decay  of  the  stored  energy  in  generator  fields.  In  these  applications  the 
conventional  rectifier  circuit  is  used  and  the  grid  firing  point  is  retarded  to  a  point  late  in 
the  positive  half  cycle.  In  the  high-power  field,  highly  specialized  inverters  have  been 
built  for  several  applications. 

IGITCTRON  RECTIFIERS.  Ignitron  rectifiers  are  ideaUy  suited  to  their  use  in  the 
high-power  field,  because  their  cathodes — a  mercury  pool — can  supply  electron  emission 
for  tremendous  overloads.  The  maximum  current  is  limited  by  the  mechanical  forces  in- 
volved and  by  the  ability  of  the  tube  to  deionize  rapidly  enough  to  prevent  arcback. 
Ignitron  rectifying  devices  are  made  in  capacities  trom  about  12  amp  up  to  1000  amp  d-c 
continuous  rating.  (In  1945  approximately  10  per  cent  of  the  central-station  power  gen- 


THYBATKON  AND  IGNITRON  CONTACTORS 


21-11 


erated  in  the  United  States  passed  through  ignitron  rectifiers.)     Continuously  evacuated 
ignitron  rectifiers  are  available  in  sizes  as  large  as  6000  kw  at  600  volts  direct. 

Figure  2  shows  the  efficiencies  of  ignitron  rectifiers  as  compared  with  synchronous  con- 
verters and  synchronous  motor-generator  sets.  The  rectifier  efficiency  increases  with 
higher  voltage  because  the  volt- 
age drop  across  the  arc  in  the 
ignitron  is  nearly  the  same  in 
all  cases,  and,  therefore,  with 
lower  d-c  voltage  the  arc  drop 
has  a  greater  proportional  effect 
than  at  high  voltage. 

Thyratron  rectifiers,  like  va- 
cuum tubes,  may  be  controlled 
with  very  minute  power  levels 
on  the  grid.  An  ignitron,  how- 
ever, requires  a  considerable 
amount  of  power  for  a  short 
time  for  firing.  This  power  may 
be  supplied  either  by  magnetic 
excitation  circuits  involving  no 
tubes  and  purely  static  control 
devices,  or  they  may  be  con- 
trolled by  electronic  firing  cir- 
cuits involving  thyratrons  in 
the  ignitor  circuit.  The  mag- 
netic firing  circuit  is  used  where 
the  maximum  of  reliability  is 
required.  Thyratron-type  firing 


93 


94 


90 


82 


I 


74 


70 


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60          80         100         120 
Per  cent  full  load,  amperes 


140         160 


FIG.  2.     Comparative  Efficiencies  of  Power-conversion  Units 


circuits,  however,  may  be  less  expensive  to  build,  and  also  their  inherently  fast  operating 
speed  may  be  more  desirable  for  some  applications. 


4.  THYRATRON  AND  IGNITRON  CONTACTORS 

Two  controlled  rectifier  tubes  of  the  thyratron  or  ignitron  type  can  be  connected  in 
inverse  parallel  and  the  combination  connected  in  series  with  a  single-phase  load  to  make 
a  single-pole  single-throw  a-c  switch  called  a  thyratron  contactor  or  an  ignitron  contactor. 
Likewise,  several  pairs  of  tubes  may  be  connected  in  polyphase  circuits  to  make  a  poly- 
phase single-throw  a-c  contactor.  This  type  of  contactor  has  several  salient  features: 

1.  In  applications  involving  a  very  large  number  of  operations,  it  eliminates  the  mechan- 
ical wear  and  resultant  maintenance  of  mechanical-type  contactors.    Furthermore,  it  is 
quiet  in  operation. 

2.  This  type  of  circuit  is  inherently  fast  in  response.     This  means  that  power  circuits 
may  be  closed  or  opened  more  rapidly  than  with  conventional  magnetic  contactors. 

3.  When  these  contactors  are  used  with  the  proper  phase  control  systems,  they  may  be 
fired  synchronously  at  a  given  phase  position  in  each  half  cycle  to  avoid  the  transient 
currents  that  will  result  in  inductive  loads  if  the  power  circuit  is  closed  at  random. 

4.  These  contactors  may  be  used  to  control  the  effective  a-c  load  voltage  by  adjustment 
of  the  firing  point.    Control  of  load  power  can  be  obtained  without  power  loss  and  without 
undue  voltage  regulation  due  to  load  changes. 

The  choice  of  thyratron  or  ignitron  tubes  is  dependent  on  the  magnitude  of  power  in- 
volved. Typical  applications  of  thyratron-type  contactors  are:  (1)  for  controlling  the 
speed  of  some  types  of  a-c  motors;  (2)  for  controlling  the  output  voltage  of  high-voltage 
transformers;  (3)  for  low-power  resistance  welders;  (4)  for  high-voltage  resistance  welders. 
By  far  the  largest  number  of  applications  of  electronic  contactors,  however,  has  been 
made  in  the  resistance  welding  field,  in  which  ignitron  tubes  are  used. 

General  requirements  for  resistance  welding  are:  (1)  single-phase  power;  (2)  power  im- 
pulses having  a  high  peak  value  over  a  short  period  of  time;  (3)  a  large  total  number  of 
impulses  over  a  given  period  of  time;  (4)  control  of  the  effective  amount  of  welding  current; 
(5)  controlled  firing  to  eliminate  transient  currents.  These  requirements  have  made  the 
ignitron  contactor  with  its  extremely  high  peak-current  capacity  and  its  controllability 
ideally  suited  to  resistance-welding  applications. 

Figure  3  illustrates  a  typical  single-phase  ignitron  contactor  for  resistance-welding  serv- 
ice. This  contactor  is  of  the  simple  on-off  type.  It  is  used  to  give  faster  operation  than 
magnetic-type  contactors  as  well  as  to  eliminate  the  maintenance  on  magnetically  operated 


21-12 


ELECTRONIC  CONTROL  EQUIPMENT 


mechanical  contactors.  Figure  4  shows  an  elementary  circuit  diagram,  for  an  on-off  type 
ignitron  contactor.  This  circuit  operates  as  follows:  When  the  initiating  switch  is  open, 
no  current  will  now  because  power  is  not  being  supplied  to  the  igniters  for  firing  the  tubes, 
Assume  now  that  the  initiating  switch  is  closed  and  that  anode  a  is  positive.  Current  will 

not  flow  through  the  left-hand  tube 
until  the  ignitor  has  been  fired. 
The  ignitor  firing  circuit  starts  with 
anode  a,  continues  through  the 
lower  right-hand  metallic  rectifier, 
through  the  water-flow  switch,  fuse,  * 
and  initiating  switch,  and  through 
the  upper  left-hand  metallic  rectifier 
into  the  ignitor  which  is  immersed 
in  the  mercury  pool.  Current  flows 
from  this  pool  to  the  primary  of  the 
welding  transformer  and  back  to 
the  other  side  of  the  a-c  line.  This 
means  that  the  entire  line  voltage 
and  the  entire  load  current  are  avail- 
able for  firing  the  ignitor.  With  this 
voltage  and  current  available,  the 
ignitor  fires  the  ignitron  tube,  and 
then  load  current  flows  through  the 
main  anode  of  the  left-hand  igni- 
tron. When  the  anode  6  is  positive, 
ignitor  firing  current  flows  through 
the  lower  left-hand  metallic  recti- 
fier, through  the  initiating  switch, 
fuse,  and  water-flow  switch;  then 
through  the  upper  right-hand  me- 
tallic rectifier  into  the  mercury  pool 
and  back  to  the  other  side  of  the  a-c 
supply.  The  metallic  rectifiers  pro- 
vide a  path  for  the  ignitor  firing 
current  during  the  half  cycle  in 
which  a  specific  ignitor  is  to  be 
fired.  During  the  other  half  cycle, 
the  metallic  rectifiers  prevent  the 
flow  of  reverse  ignitor  current  which 
would  damage  the  ignitor. 

Figure  5  shows  how  thyratron 
tubes  may  be  used  to  control  the  ignitron  power  tubes  in  order  to  obtain  phase  control 
or  synchronous  timing  control  of  the  output  power.  A  resistor  is  used  in  the  anode  of 
the  two  thyratron  tubes  to  limit  the  peak  current  flowing  while  a  fuse  is  used  to  protect 
the  tubes  against  high  rms  current  which  would  result  if  the  ignitor  were  continuously  fired 
with  no  load  current  flowing. 


FIG.  3.    Single-phase  Ignitron  Contactor  with  Size  D  Tubes 
(Courtesy  General  Electric  Co.) 


Welding  <   (       1 

transformed    v» J 

f     Work 


Fro.  4.    Circuit  Diagram  for  an  Ignitron  Contactor  with  Metallic  Rectifier  Firing 


Figure  6  is  a  typical  duty  cycle  rating  curve  for  four  sizes  of  contactors  (ratings  are  for 
two  tubes)  used  on  a  460-volt  power  supply.  Three  factors  affect  the  amount  of  current 
which  can  be  handled  by  a  pair  of  ignitrons:  (1)  The  maximum  current  is  a  function  of  the 
line  voltage;  for  low  line  voltages,  the  tubes  will  carry  a  higher  current  than  for  high  line 


FUNDAMENTAL   ELECTRONIC   CONTROL   CIRCUITS      21-13 

voltages.  (2)  For  a  given  line  voltage  there  is  a  maximum  rms  current  rating  irrespective 
of  the  length  of  time  the  current  flows.  (3)  If  the  current  flows  over  an  appreciable  length 
of  time,  even  though  the  duty  cycle  is  low,  the  averaging  time  of  the  tube  must  be  con- 
sidered to  prevent  overloading  the  tube  from  a  thermal  standpoint.  Per  cent  duty  shown 
in  Fig. '  6  indicates  the  percentage  of  total  time  that  current  is  flowing  through  the  tubes. 


FIG.  5.     Ignitron  Contactor  with  Thyratron  Firing  Circuit 

A  pair  of  size  D  tubes  has  a  continuous  rating  of  800  amp.  As  the  duty  cycle  is  reduced, 
however,  the  maximum  current  can  be  increased.  If  the  time  involved  is  short,  the  rating 
can  reach  almost  5000  amp.  The  size  D  tubes  have  an  averaging  time  of  5.6  sec.  This 
means  that,  for  any  5.6-sec  period,  the  average  current  should  not  exceed  800  amp  rms, 
even  though  the  actual  current  during  conduction  equaled  5000  anip.  In  other  words, 
the  tubes  could  carry  5000  amp  for  approximately  0.9  sec  (800/5000  X  5.6),  provided  no 


D  to  50C 

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11  se 
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Per  cent  doty 
FIG.  6.     Ignition-contactor  Duty-cycle  Rating  Curve 

current  was  carried  during  the  remainder  of  the  5.6-sec  period.  If  it  were  necessary  to 
have  current  flowing  over  a  5.6-sec  period  continuously,  the  tube  rating  would  be  the  same 
as  for  100  per  cent  duty,  or  800  amp. 


FUNDAMENTAL  ELECTRONIC  CONTROL  CIRCUITS 

Because  the  applications  of  electronic  control  are  so  diversified  and  because  most  appli- 
cations are  highly  specialized  it  is  impractical  to  describe  even  a  moderately  complex  circuit 
with  all  its  ramifications.  An  attempt  will  be  made  in  this  section  to  discuss  several 
fundamental  control  circuits  which  are  commonly  used  in  complete  electronic  systems, 
with  the  hope  that  these  circuits  will  be  recognized  when  they  are  a  part  of  complete 
systems. 


21-14 


ELECTRONIC   CONTROL  EQUIPMENT 


5.  STABILIZED  D-C  CONTROL  POWER  SUPPLIES 

Many  electronic  control  equipments  include  a  small  d-c  power  supply  usually  consisting 
of  a  diametric  (full-wave)  rectifier  and  a  suitable  filter.  These  are  similar  to  the  power 
supplies  used  in  radio  receivers  which  are  discussed  in  Section  7, 

In  order  to  provide  stability  of  operation  of  the  associated  electronic  control  equipment, 
it  is  often  necessary  to  provide  a  d-c  output  voltage  which  is  held  constant  irrespective  of 
line  voltage  or  changes  in  load.  This  may  be  accomplished  in  several  ways;  the  most 
common  are: 

1.  A-c  voltage  stabilizers.  Figure  1  illustrates  one  typical  type  of  automatic  voltage 
stabilizer  circuit.  The  voltage-regulating  action  is  due  primarily  to  reactor  1  and  the 
parallel  capacitor.  When  the  line  voltage  is  high,  reactor  1,  which  operates  near  the 
knee  of  the  B-H  curve,  becomes  saturated.  Under  this  condition  the  reactor  current 
and  the  capacitor  current  are  about  equal,  the  total  current  being  at  about  unity  power 
factor.  As  a  result  there  is  a  voltage  drop  across  reactor  2.  When  the  line  voltage  drops, 


Reactor  2 


Variable 

Input 
voltage 


Reactor  1  ; 


Constant 
output 
voltage 


Capacitor 


+  ' 

rh      Series 
M  regulating 

resistor 

D-c  o 

utput                   _ 

+ 

from  filter                 /"J 

—  \  Regulating 

V 

'•/       glow 

V. 

S       tube 

Regulated 
d-c  voltage 


FIG.  1.     Circuit  Diagram  of  A-c  Voltage  Stabilizer       FIG.  2.    Glow-tube-type  D-c  Voltage  Regulator 

however,  the  reactor  current  falls  off  more  rapidly  than  the  capacitor  current.  As  a 
result  the  current  through  reactor  2  is  predominantly  capacitive  and  there  is  a  rise  in 
voltage  across  reactor  2.  Units  of  this  type  will  hold  the  output  voltage  to  within  =bl 
per  cent  over  about  a  ±15  per  cent  change  in  input  voltage. 

2.  Gaseous-type  voltage-regulating  glow  tubes.    These  tubes  have  inherently  a  constant 
voltage  drop.    When  the  voltage  has  reached  the  ionization  point  a  very  small  increase 
in  applied  voltage  will  result  in  a  comparatively  large  increase  in  tube  current.    These 
tubes  are  well  suited  to  control  power  circuits  involving  small  currents.     Three  ratings 
are  most  commonly  used,  namely,  75,  105,  and  150  volts  at  40  ma  maximum.    Figure  2 
shows  a  regulating  circuit,  using  a  glow  tube.    The  constant-voltage  characteristic  of  the 
glow  tube  will  permit  the  rectifier  output  voltage  to  be  increased  or  decreased,  and  the 
voltage  difference  will  be  largely  absorbed  by  the  series  regulating  resistor.    This  results 
in  a  nearly  constant  output  voltage  across  the  glow  tube. 

3.  A  series-type  voltage  regulator.    This  regulator  uses  a  glow  tube  merely  as  a  reference 
voltage  and  involves  a  regulating  principle  which  will  be  discussed  later.    This  type  of 
voltage-regulating  system  will  provide  larger  amounts  of  voltage  and  current  and  at  greater 
accuracy  than  can  generally  be  obtained  with  the  simple  glow-tube  arrangement  discussed 
before. 

6.  TIMING  CIRCUITS 

Timing  circuits  are  widely  used  in  electronic  control  systems.  In  many  equipments 
such  as  general-purpose  timers,  resistance-welder  sequence  timers,  timers  for  cut-off 
register  applications  which  operate  correcting  devices  for  a  given  length  of  time,  and 
timing  circuits  which  control  the  acceleration  or  deceleration  rate  of  motors,  timing  is  a 
definite  function.  Often,  however,  the  use  of  timing  circuits  may  be  incidental  to  the 
function  to  be  performed  by  a  given  equipment.  For  example,  filter  capacitors  in  d-c 
control  power  supplies,  capacitors  across  relay  coils  which  are  operated  by  half-wave 
rectifiers,  resistor-capacitor  combinations,  and  electronic  relays  involving  impulses  of  very 
short  duration  which  are  used  to  maintain  an  operating  signal  long  enough  to  operate  a 
magnetic  relay  are  all  energy-storage-type  timing  circuits. 

A  capacitor  is  used  as  a  basic  timing  element  in  most  timing  circuits.  Figure  3A  shows 
a  simple  series  resistor-capacitor  circuit.  When  the  switch  is  closed,  the  capacitor  charges 
through  the  series  resistor.  The  instantaneous  voltage  across  the  capacitor  is  expressed 


TIMING  CIRCUITS 


21-15 


Time 


Switch 
clcsei 


Switch 


<U        L/opened 


Time 


FIG.  3.     Basic  R-C  Timing  Circuits 


by  the  equation  ect  =  E(l  —  e  */rc),  where  t  =  time  in  seconds  after  the  switch  is  closed. 
Figure  3B  shows  a  similar  circuit  in  which  the  capacitor  is  charged  almost  instantly 
(assuming  that  the  d-c  source  has  no  resistance)  and  in  which  the  capacitor  discharges 
through  the  parallel  resistor.  The  instantaneous  capacitor  voltage  after  opening  the 
switch  is  expressed  by  the  equa- 
tion ect  =  E  X  e~i/ro.  The  above 
equations  will  give  the  voltage 
at  any  time.  A  more  generally 
used  term,  however,  is  the  time 
constant.  The  time  constant  is 
defined  as  the  time  at  which  the 
capacitor  has  charged  to  approxi- 
mately two-thirds  of  its  final 
voltage  value  or  the  time  at 
which  the  capacitor  has  dis- 
charged to  approximately  one- 
third  of  its  initial  voltage  value. 
The  time  constant  is  expressed 
as  T  —  rCj  where  T  =  time  in 
seconds,  r  —  resistance  in  ohms, 
and  C  =  capacitance  in  farads^. 
Often  a  basic  timing  circuit  will 
include  a  parallel  r-C  circuit 
with  a  resistor  in  series.  This 
will  result  in  a  time  lag  in  the 
charging  circuit  and  a  lag  in  dis- 
charging as  well, 

r-C  circuits  are  used  in  complete  circuits  to  obtain  a  time  delay  by  applying  the  capacitor 
voltage  to  a  tube  grid  to  render  the  tube  conducting  or  non-conducting  at  a  given  capacitor 
voltage.  Although  different  detail  circuit  arrangements  are  used  in  timing,  the  basic  cir- 
cuits shown  are  common  to  most  electronic  timers. 

A  general-purpose  time-delay-relay  circuit  which  uses  a  parallel  r-C  circuit  for  timing 
is  shown  in  Fig.  4.  This  is  typical  of  complete  timing  circuits.  The  coil  of  the  relay  is 
energized  a  definite  time  after  switch  S  is  closed.  When  switch  S  is  open,  the  cathode  of 
the  vacuum  tube  is  connected  through  2R  (which  is  a  relatively  low  resistance)  to  the 
anode.  Resistor  1R  and  the  potentiometer  P  form  a  voltage  divider  across  the  a-c  power 
supply.  With  the  switch  open,  the  cathode  is  effectively  connected  to  line  1  (through 
resistor  2R) .  Therefore  current  will  now  from  line  3  through  a  section  of  the  potentiometer, 
through  the  parallel  resistor-capacitor  combination  into  the  grid,  and  back  through  the 
cathode.  The  grid  in  this  instance  acts  as  an  anode  since  it  is  positive  with  respect  to  the 
cathode  during  every  other  half  cycle.  The  voltage  drop  across  the  timing  resistor  R 
causes  capacitor  C  to  become  charged.  As  capacitor  C  charges,  the  voltage  applied  to 
the  grid  during  alternate  half  cycles  becomes  less  positive — the  grid  voltage  is  the  alge- 
braic sum  of  the  a-c  voltage  and  the 
capacitor  voltage.  The  capacitor  will  con- 
tinue to  charge  until  the  voltage  across  the 
capacitor  is  equal  to  the  peak  of  the  a-c 
voltage  from  line  1  to  the  slider  of  the 
potentiometer.  If  the  potentiometer  P  is 
turned  to  the  extreme  counterclockwise 
position,  this  will  be  the  peak  line  volt- 
age. As  potentiometer  P  is  turned  clock- 
wise, the  capacitor  charges  to  a  lower 
value.  When  switch  5  is  closed  the 
cathode  is  connected  to  the  other  side  of 
the  line.  At  this  instant  the  grid  of  the 
tube  is  negative  with  respect  to  the  cath- 
ode by  whatever  potential  the  capacitor  C 
is  charged.  The  instantaneous  grid  volt- 
age, however,  is  a  summation  of  the  d-c  capacitor  voltage  and  the  a-c  voltage  from  line  3 
to  the  potentiometer  slider.  Figure  5  shows  the  action  of  the  capacitor  discharge  circuit 
plus  the  a-c  component.  It  can  be  seen  that,  when  the  switch  is  first  closed,  the  grid  is 
sufficiently  negative  at  all  times  so  that  no  plate  current  flows.  As  the  capacitor  dis- 
charges, however,  the  grid  potential  reaches  a  point  where  sufficient  plate  current  flows  to 
energize  the  relay  in  the  plate  circuit.  The  capacitor  C\  of  Fig.  4  across  the  relay  coil  is 


3  S  6 

FIG.  4.     General-purpose  Time-delay-relay  Circuit 


21-16 


ELECTRONIC   CONTROL  EQUIPMENT 


used  as  an  energy-storage  device  during  the  half  cycle  in  which  the  tube  is  conducting  so 
that  it  can  supply  energy  to  the  coil  during  the  half  cycle  when  the  tube  is  not  conduct- 
ing, thus  preventing  the  relay  from  chattering. 


irid  voltage 


Timing 
capacitor 
voltage 


A     Tube 

MM  current 


Switch 
dosed 


Time. 


FIG.  5.    Grid  Voltage  and  Anode  Current  after 
Timing  Switch  of  Timer  (Fig.  4)  is  Closed 


>  d 

?SR 

*     + 

J 

( 

L^ 

Tube  2 

5> 

D-c 


r 

[ 

-SUf 

piy 

Tube  V- 

J 

SIgnaJ 

r 

N 

.— 

FIG.  6.     Typical  D-c  Voltage  Amplifier 


7.  D-C  AMPLIFIERS 

D-c  amplifiers  are  commonly  used  in  photoelectric  controls,  motor  and  generator  con- 
trols, electronic  voltage  regulators,  regulated  battery  chargers,  and  other  equipment 
where  it  is  desirable  to  amplify  d-c  signals.  These  amplifiers  are  generally  not  too  well 
suited  for  applications  where  the  signal  voltage  is  low  and  where  a  high  degree  of  ampli- 
fication is  needed,  because  the  instability  of  the  tube  characteristics  can  result  in  output 
variations  equal  to  or  greater  in  magnitude  than  would  be  obtained  from  a  complete  input- 
signal  range.  D-c  amplifiers  are  very  well  suited,  however,  to  regulating  circuits  where  the 
difference  between  two  fairly  large  voltages  is  applied  to  the  grid  circuit.  Regulating 
circuits  will  be  discussed  in  more  detail  later. 

Figure  6  shows  a  typical  d-c  voltage  amplifier.  Resistors  IE  and  2R  form  a  voltage 
divider  which  establishes  the  cathode  potential  of  tube  2.  Resistors  3/jJ,  4R,  and  5R  are 
chosen  so  that,  when  plate  current  is  not  flowing  in  tube  1,  the  grid  of  tube  2  is  positive. 
When  current  flows  through  tube  1,  the  additional  IR  drop  in  resistor  3R  lowers  the  grid 
voltage  of  tube  2  and  reduces  its  plate  current.  In  the  plate  circuit  of  tube  2  is  shown  a 
saturable  reactor  which  could  be  used  in  a  thyratron  phase-shifting  circuit,  although  any 
voltage-  or  current-responsive  device  could  also  be  used  here.  If  the  signal  voltage  were 
zero,  the  grid  of  tube  1  would  be  at  cathode  potential  and  tube  1  would  be  carrying  approx- 
imately maximum  current.  As  the  grid  voltage  of  tube  1  is  made  negative,  the  plate 
current  of  tube  1  would  decrease,  thus  increasing  the  plate  current  in  tube  2. 


8.  A-C  AMPLIFIERS 

In  control  circuits  a-c  amplifiers  are  usually  of  the  capacitance-coupled  type.  Though 
in  some  circuits  they  are  used  in  much  the  same  way  as  in  radio  circuits  they  are  often 
employed  in  circuits  in  which  it  is  desirable  to  amplify  an  impulse  signal.  For  example, 
in  photoelectric  cutoff  or  web-register  control  systems  used  in  cutting  bags  or  labels  at  a 
particular  point  with  respect  to  the  printed  material  it  is  desirable  for  the  equipment  to 
respond  to  small  marks  on  paper  which  is  moving  at  high  speed.  The  light  impulse  is 
obtained  by  scanning  the  paper  surface  with  a  photo  tube.  The  normal  light  level  may 
be  relatively  high,  and  the  mark  on  the  paper  may  correspond  to  only  a  small  change  in 
light.  Therefore,  it  is  necessary  to  have  a  very  sensitive  amplifier  which  responds  only 
to  rapid  light  impulses  or  changes  and  not  to  steady-state  light  or  general  changes  in 
light  level. 


REGULATING  CIRCUITS 


21-17 


7R 


2C            . 

M,       ' 

INTube  1 

r 

V 

I  Relay 


sjube  2 


A  typical  impulse  amplifier  is  shown  in  Fig.  7.  Resistor  IR,  2R,  3R,  and  4R  constitute 
a  voltage  divider  to  supply  the  various^voltage  levels  required  in  the  circuit.  The  proper 
grid  bias  on  tubes  1  and  2  is  supplied  through  grid  resistors  6R  and  8R  respectively. 
Assume  that  a  steady  level  of  light  is 

applied  to  the  phototube.    The  circuit  Reset 

constants  are  arranged  so  that  under 
this  condition  current  flows  in  tube  1 
but  not  in  tube  2.  Now  if  light  sud- 
denly is  diminished  the  phototube 
current  is  reduced  and  capacitor  1C 
momentarily  "pulls"  the  grid  of  tube 
1  to  a  more  negative  position.  This 
immediately  reduces  the  plate  current 
in  tube  1,  which  in  turn  raises  mo- 
mentarily the  grid  voltage  of  tube  2 
through  capacitor  2(7.  Tube  2  is  a 
thyratron  type  and  has  a  d-c  anode 
supply  voltage.  Therefore  although 
its  grid  voltage  may  go  in  the  positive 
direction  only  momentarily  this  is 
sufficient  to  fire  the  tube  and  once 
fired  it  remains  conducting  until  the 
reset  contact  shown  is  opened. 


D8R 


FlQ.  7.   Typical  Impldse.type  A.c  Amplifier  M  Used 
Photoelectric  Cutoff  Register  Controls 


9.  REGULATING  CIRCUITS 

Regulating  circuits  are  used  in  many  applications  where  it  is  desirable  to  hold  quantities 
such  as  voltage,  current,  speed,  position,  pressure,  temperature,  and  the  like  constant 
irrespective  of  conditions,  such  as  load  or  line-voltage  variations,  which  would  normally 
cause  the  quantity  being  held  to  deviate  from  the  desired  value.  Any  regulating  circuit 
has  four  fundamental  requirements : 

1.  A  standard  or  a  reference  voltage,  which  is  held  constant  at  all  times  and  which 
represents  a  value  against  which  the  quantity  to  be  held  constant  is  compared.    The  refer- 
ence voltage  is  the  heart  of  a  regulating  circuit  because  the  accuracy  of  the  regulator  can 
never  be  more  accurate  than  the  reference  voltage. 

2.  Means  must  be  provided  so  that  the  quantity  to  be  regulated  will  produce  a  signal 
voltage  which  can  be  compared  with  the  reference  voltage. 

3.  An  amplifier,  to  amplify  any  difference  that  may  exist  between  the  regulated]quantity 
and  the  reference  voltage. 

4.  A  controlling  means,  which  will  operate  from  the  amplifier  to  restore  the  quantity 
to  approximately  the  yalue  originally  held  before  the  deviation  occurred. 

Figure  8  shows  an  electronic  voltage  regulator,  the  purpose  of  which  is  ,to  supply  a 
constant  d-c  voltage  output  over  a  wide  range  of  load  current  and  d-c  input  line  voltage. 
The  total  output  power  from  this  circuit  is  limited  by  the  current  that  can  be  handled 
by  the  2A3  tube  shown.  (Parallel  tubes  or  other  tube  types  will  deliver  more  power.) 
Assume  for  the  moment  that  switch  S  is  open  and  also  that  no  current  is  flowing  in  the 
plate  of  the  6J7  tube.  Under  these  conditions,  the  grid  of  the  2A3  tube  will  be  at  approxi- 
mately cathode  potential.  The  effective  resistance  of  the  2A3  tube  under  this  condition 
is  very  low.  Therefore  a  high  voltage  will  exist  across  lines  2  and  3.  Resistors  IR  and  2Rt 
which  are  in  series  with  the  OA3/VR75  glow  tube,  are  such  that  with  rated  regulated  d-c 
output  voltage  the  glow  tube  will  ionize  and,  once  ionized,  will  conduct  a  small  amount  of 
current.  The  characteristic  of  the  glow  tube  is  such  that,  if  the  voltage  between  lines  2 
and  3  is  varied,  the  voltage  across  the  glow  tube  will  remain  reasonably  constant.  How- 
ever, the  more  constant  the  voltage  across  lines  2  and  3  can  be  held,  the  less  variation 
there  will  be  in  the  voltage  across  the  glow  tube  because  the  voltage  across  the  glow  tube 
changes  somewhat  with  current.  Since  the  objective  of  this  equipment  is  to  hold  the 
voltage  output  constant,  the  reference  voltage  must  be  held  as  constant  as  possible. 

The  cathode  of  the  6J7  amplifier  tube  is  connected  to  the  glow-tube  anode,  point  4. 
Resistors  3R  and  4#  constitute  a  voltage  divider  across  the  output  of  the  regulator  and 
therefore  provide  a  signal  to  the  grid  of  the  6J7  tube  which  is  proportional  to  the  output 
voltage.  Resistors  3JK  and  4JS  are  selected  so  that  with  rated  regulated  d-c  output  voltage 
the  grid  of  the  6J7  tube  will  be  at  approximately  cutoff.  The  actual  grid  signal  consists 
of  the  difference  between  the  voltage  across  resistor  4.R  and  the  voltage  across  the  glow 
tube.  If  it  is  assumed  that  the  reference  voltage  from  the  glow  tube  is  constant,  then 


21-18 


ELECTRONIC  CONTROL  EQUIPMENT 


any  deviation  in  output  voltage  will  be  reflected  as  a  difference  in  the  potential  between  the 
grid  and  the  cathode  of  the  6J7  tube. 

As  was  stated  before,  if  no  current  is  flowing  in  the  6J7  tube,  the  effective  resistance 
of  the  2A3  tube  is  very  low.     The  output  voltage  between  lines  2  and  3  therefore  will  be 


Load 


FIG.  8.     Electronic  D-c  Voltage  Regulator 

high.  This,  however,  results  in  a  grid-to-cathode  voltage  on  the  6J7  tube  of  a  sufficiently 
high  positive  value  to  cause  the  6J7  to  conduct  plate  current.  This  current  causes  an 
IR  drop  between  lines  2  and  7  across  resistor  5R,  and  causes  the  grid  of  the  2A3  tube  to 
assume  a  negative  potential.  This  negative  potential  increases  the  effective  resistance 
of  the  2A3  tube  and  thus  reduces  the  output  voltage  between  lines  2  and  3.  There  is  a 

certain  grid-to-cathode  voltage  on 
the  2A3  tube  which  will  result  in  an 
output  voltage  across  lines  2  and  3 
sufficient  to  adjust  the  grid  potential 
of  the  6J7  tube  to  give  the  desired 
grid-to-cathode  voltage  on  the  2A3 
tube. 

If  now  the  regulator  is  holding  a 
given  voltage  and  load  is  added  by 
closing  the  switch  S,  the  output  volt- 
age between  lines  2  and  3  will  im- 
mediately drop  because  the  effective 
resistance  of  the  2A3  tube  is  in  series 
with  the  load.  In  order  to  correct 
this  low-voltage  condition,  the  regu- 
lator must  again  go  into  action. 
When  the  output  voltage  is  reduced, 
the  grid  voltage  is  made  more  nega- 
tive with  respect  to  the  cathode 
voltage  on  the  6J7  tube.  This  more 
negative  value  of  grid  voltage  re- 
duces the  6J7  plate  current  and 
thereby  makes  the  2A3  grid  less  neg- 
ative, which  reduces  the  equivalent 
resistance  of  the  2A3  tube,  again  re- 
storing the  correct  output  voltage. 
The  curves  of  Fig.  9  show  input 


^ 

X, 

""** 

X 

X 

xj{ 

put 

volt 

300 

"^ 

X, 

*x 

». 

120 

^X 

"X 

x 

^ 

"*** 

100 

±5 
° 

\ 

6J 

7  pic 
nicrc 

te  c 
)am 

urre 
Dere 

nt 

;) 

0 

•  250 

1 

80 

X 

V. 

s^ 

60 

X 

X 

s 

40 

200 

Re 

jula 

ed  i 

5Utp 

Jt  V 

alts 

N 

20 

170 

Q 

) 

1 

0 

2 

0 

3 

0 

FIG.  9. 


Performance  of  Electronic  Voltage  Regulator 
Shown  in  Fig.  8 


Load  mtlliamperes 

and  output  voltage  as  a  function  of 
load  current  for  the  regulator.  It 
also  shows  the  plate-current  change 
required  of  the  6J7  tube  to  give  accurate  regulation  of  the  output  voltage.  It  can  be  seen 
that,  for  a  wide  change  in  output  load  current  and  for  a  wide  change  in  input  line  voltage, 
the  regulated  output  voltage  is  held  nearly  constant. 

The  electronic  voltage  regulator  just  discussed  is  a  simple  unit  when  compared  with  most 


THYRATRON  GRID-CONTROL  CIRCUITS 


21-19 


complex  electromechanical  regulators;  however,  it  is  typical  of  all  regulating  circuits  in 
that  it  includes  the  fundamental  elements  of  a  regulating  circuit. 

The  essential  regulating  elements  can  be  explained  as  follows:  Referring  to  Fig,  8,  the 
voltage  across  the  glow  tube  between  lines  3  and  4  is  the  standard  or  reference  voltage. 
The  voltage  across  lines  3  and  6  is  the  signal  voltage.  Any  difference  between  the  voltages 
4  to  3  and  6  to  3  is  applied  to  the  grid  of  the  6J7  tube,  which  amplifies  the  existing  devia- 
tion. This  in  turn  controls  the  grid  on  the  2A3  tube  to  correct  for  the  voltage  deviation. 
An  important  feature  of  a  regulating  circuit,  as  far  as  obtaining  accuracy  is  concerned,  is 
the  use  of  two  relatively  large  voltages  in  the  comparison  circuit.  This  can  be  shown  best 
by  an  example.  If  the  6J7  tube  required  0.1-volt  change  on  the  grid  to  effect  a  complete 
swing  of  the  grid  of  the  2 A3  tube,  this  0.1-volt  change  can  be  accomplished  on  a  200-volt 
output  circuit  with  a  75-volt  reference  voltage  by  approximately  a  0.3-volt  total  error 
in  the  output.  In  other  words,  the  inherent  regulating  error  of  the  system  does  not  exceed 
0.3  volt  out  of  200  volts  or  0.15  per  cent  for  a  wide  load  and  input  line-voltage  change. 
Furthermore,  if  the  inherent  tube  characteristics  varied  in  such  a  manner  that  a  given 
6J7  plate  current  resulted  with  a  grid  voltage  0.3  volt  different  from  the  original  voltage 
the  output  voltage  would  again  be  in  error  by  only  approximately  0.15  per  cent.  If  lower 
voltages  were  used,  both  errors  would  increase. 


10.  THYRATRON  GRID-CONTROL  CIRCUITS 

Thyratron  tubes  are  used  in  circuits  having  d-c  or  a-c  anode  supplies.    When  they  are 
used  with  d-c  supplies  they  are  on-off  devices  and  a  simple  d-c  grid-bias  control  can  be 
used  to  fire  the  tubes.    In  a-c  cir- 
cuits, such  as  rectifiers  or  electronic       .  ^Anode  voltage 
contactors,  however,  it  is  usually 
desirable  to  control  the  effective 
output  voltage  by  controlling  the 
point  in  each  cycle  at  which  a  par-      0 
ticular  tube  fires. 

When  a  thyratron  is  connected 
to  an  a-c  power  supply  the  anode 
voltage  is  different  at  each  point 
in  the  cycle.  Therefore  at  the  be- 
ginning of  a  positive-voltage  half 
cycle  a  positive  value  of  grid  volt- 
age is  required  to  fire  the  tube.  As 
the  anode  voltage  increases  in  the  positive  direction,  the  grid  voltage  required  to  fire  the 
tube  becomes  negative.  Figure  10  shows  a  curve  of  the  grid  voltage — the  critical  grid- 
voltage  curve — required  to  fire  a  thyratron  tube  on  an  a-c  power  supply.  Figure  1QA 
shows  that  with  an  a-c.  grid  voltage  applied  180°  out  of  phase  with  the  anode  voltage  the 
tube  does  not  conduct.  Figure  10B  shows  that  with  an  a-c  grid  voltage  applied  in  phase 
with  the  anode  voltage  the  tube  conducts  for  the  entire  half  cycle.  Figure  IOC  shows 
that  with  the  a-c  grid  voltage  lagging  the  applied  voltage  by  90°  the  tube  conducts  for 
half  of  the  half  cycle. 

Although  a  number  of  different  types  of  grid-controlled  circuits  can  be  used  for  con- 
trolling the  output  voltage  of  thyratron  tubes,  the  most  widely  used  control  circuits  are: 
(1)  grid-voltage  phase  shifting;  (2)  fixed  90°  phase  shift  with  adjustable  d-c  bias. 


Critical 

grid 
voltage 

(A)  (B)  (C) 

Zero  output  Full  output  Half  output 

voltage  voltage  voltage 

FIG.  10.     Method  of  Grid  Contiol  of  Thyratron  Tubes 


FIG.  11.     Basic  Circuit  for  Controlling  Thyratron  Tubes  with  Adjustable  Grid-voltage  Phase  Angle 

Figure  1 1  shows  the  basic  circuit  for  a  method  of  controlling  thyratrons  by  means  of  an 
a-c  grid  potential  whose  phase  angle  can  be  adjusted.    The  grid-voltage  phase  position  is 


21-20 


ELECTRONIC  CONTROL  EQUIPMENT 


determined  by  the  ratio  between  the  variable  inductance  and  the  fixed  resistance  shown 
in  the  primary  of  the  grid  transformer  2T.  Figure  12  shows  the  vectors  of  the  voltages 
involved  in  the  grid  circuit.  EI  is  the  anode-transformer  secondary  voltage.  Ir  is  the 
voltage  across  the  resistor  and  is  in  phase  with  the  resistor-reactor  series  current.  !XL  is 

the  voltage  across  the  variable  reactor.  The  actual  grid 
voltage  Eg  can  be  seen  to  vary  in  phase  position,  but 
not  in  magnitude,  as  IXL  is  changed  (change  in  the  re- 
actance). The  variable  reactor  could  be  an  iron-core 
reactor  with  a  removable  plunger,  or  a  small  saturable 
reactor,  the  d-c  winding  of  which  could  be  energized  by 
a  vacuum  tube  or  other  means.  It  is  also  possible  to 
use  a  fixed  capacitor  and  a  variable  resistor  to  obtain 
this  same  type  of  control.  Another  method  of  obtain- 
ing this  same  type  of  control  is  by  the  use  of  a  selsyn 
(induction  phase  shifter)  which  has  a  three-phase 
primary  stator  and  a  single-phase  secondary  rotor.  As 
the  rotor  is  turned,  the  rotor  output  voltage,  which  can  be  applied  directly  to  the  grid 
circuit,  changes  in  phase  position. 

Figure  13  shows  the  basic  circuit  for  obtaining  a  fixed  a-c  voltage  phase  shift  plus  an 
adjustable  d-c  voltage  for  controlling  the  thyratron  firing  point.  The  voltage  Eg  can  be 
seen  to  be  made  up  of  two  components:  Ez,  having  a  magnitude  which  is  determined  by 
the  ratio  of  transformer  2T,  and  a  phase  position  which  is  determined  by  the  ratio  of 
resistance  and  capacitance  in  the  primary  of  transformer  2T;  and  an  adjustable  d-c 


PIG.  12.    Vector  Voltages  of  Phase- 
shift  Circuit  of  Fig.  11 


90°  lagging 
a-c  grid  potential,  Eg 

FIG.  13.    Basic  Circuit  for  Controlling  Thyratron  Tubes  with  Fixed  A-c  Voltage  Phase  Shift  plus  Ad- 
justable D-c  Grid  Voltage 

potential  EI.  With  this  circuit,  grid  control  is  obtained  by  adjusting  the  d-c  grid  poten- 
tial EI.  Figure  13  shows  firing  full  on  with  positive  grid  voltage  and  nearly  full  off  with 
negative  grid  voltage.  Smooth  control  can  be  obtained  over  the  entire  range. 


COMPLETE  ELECTRONIC  DEVICES 

Many  electronic  circuits  can  be  built  into  complete  electronic  devices,  the  operation  of 
which  is  independent  of  associated  equipment.  Timing  relays,  electronically  regulated 
power  supplies,  rectifiers,  and  electronic  contactors  are  complete  electronic  devices  which 
have  been  discussed  previously. 


11.  ELECTRONIC  RELAYS 

CONTACT-OPERATED  ELECTRONIC  RELAYS.  Contact-operated  electronic  relays 
are  used  where  it  is  desirable  to  operate  a  magnetic  contactor  or  other  electrically  operated 
device  upon  the  closure  of  a  circuit  which  has  insufficient  current-carrying  capacity  to 
operate  the  final  or  an  intermediate  device.  For  example,  it  is  often  desirable  to  have  a 
power  circuit  initiated  when  an  instrument  pointer  reaches  a  certain  mark  on  the  scale. 
The  pointer  contacts  in  most  cases  are  not  only  inadequate  to  carry  the  power  in  the  load 
circuit  but  also  inadequate  to  carry  the  power  required  to  actuate  a  contactor  in  the  load 
circuit.  The  contacts,  however,  are  adequate  for  insertion  in  the  grid  circuit  of  an  elec- 
tronic relay,  which  will  in  turn  initiate  a  magnetic  contactor.  Other  applications  of  this 
type  of  relay  are:  (1)  liquid-level  controls  where  the  liquid  itself  is  the  conducting  me- 


ELECTRONIC  RELAYS 


21-21 


Typical  Circuit  Diagram   for  Contact- 
operated  Electronic  Relay 


dium — in  this  case  a  pump  or  a  valve  is  operated  by  means  of  the  relay;  (2)  high-low 
gaging  _  of  small  parts  where  the  parts  passing  between  contacts  carry  the  grid  signal 
which  in  turn  operates  the  relay  to  actuate  reject  devices;  (3)  drop-switch  circuits  in 
textile  mills  to  operate  signals  which  indicate  broken  "ends"  of  yarn. 

Figure  1  shows  a  typical  circuit  diagram 
for  a  contact-operated  electronic  relay. 
When  the  switch  S  is  open,  the  transformer 
secondary  supplies  power  through  the  half- 
wave  rectifier  to  charge  capacitor  2C.  At 
the  same  time  the  grid  of  the  amplifier  tube 
is  at  cathode  potential.  The  triode  tube 
conducts  full  current  and  relay  CR  is  en- 
ergized. When  the  switch  S  is  closed,  the 
grid  of  the  amplifier  tube  is  brought  to  a 
negative  potential  and  the  relay  CR  is  de- 
energized.  By  the  proper  choice  of  resis- 
tors 1C,  IR,  and  2R,  this  circuit  will  operate 
with  resistances  as  high  as  10  megohms 
across  the  switch  terminals.  Also,  with 
other  resistor  and  capacitor  values  it  will 
operate  when  the  switch  circuit  is  closed  yIG> 
only  long  enough  to  give  a  very  short  im- 
pulse for  charging  capacitor  1C. 

PHOTOELECTRIC  RELAYS.  Photoelectric  relays  are  often  used  as  limit  switches 
for  operating  signals,  counters,  or  other  devices  in  applications  where  mechanical  limit 
switches  would  be  unsuitable  because  of  an  excessive  number  of  operations,  because  of 
extremely  high  velocities  of  moving  articles,  because  of  temperature  extremes  or  because 
the  material  which  is  to  operate  the  device  has  insufficient  mechanical  strength  to  operate 
a  mechanical  limit  switch.  There  are  also  many  photoelectric-relay  applications  which 
are  unique  to  light-responsive  devices,  as  indicated  by  the  following  examples:  (1)  oper- 
ation by  the  light  reflected  by  or  transmitted  through  certain  colors  of  material;  (2)  oper- 
ation by  the  amount  of  light 
transmitted  through  holes  in 
cloth,  paper,  steel,  or  other 
material;  (3)  operation  by 
light  reflected  from  certain 
types  of  bottle  cracks. 

Figure  2  illustrates  an  in- 
genious application  of  a  sim- 
ple photoelectric  relay.  A 
motor-driven  screwdriver  is 
used  to  make  an  adjustment 
on  a  relay  contact.  When 
the  contact  opens,  an  indicat- 
ing light  is  turned  on.  A 
phototube  is  held  over  the 
indicating  light  to  shut  off 
the  motor-driven  screwdriver 
automatically  when  the  light 
comes  on,  thus  assuring 
proper  adjustment  of  the 
relay. 

A  number  of  special  and 
general-purpose  types  of 


photoelectric  relays  are  com- 
mercially available.    FigureS 
shows  several  general-purpose 
photoelectric  relays:  1  is  an 
FIG.  2.     Application  of   Photoelectric  Relay  to  Motor-operated    out(jOOr-type      weatherproof 
Screwdriver  (Courtesy  General  Electric  Co.)  relay*    2    is    an    indoor-type 

high-sensitivity  relay  with  a  large  light-collecting  lens;  3  is  a  high-sensitivity  relay  with  a 
separately  mounted  phototube;  4  is  a  simple  general-purpose  relay;  5  is  a  general-purpose 
relay  with  a  separately  mounted  phototube.  w 

Figure  4  shows  the  circuit  diagram  for  the  general-purpose  relay,  numbered  4  in  Fig.  3. 
Assume  that  light  is  not  shining  on  the  phototube  and  that  the  slider  of  potentiometer  P 


\ 


21-22 


ELECTRONIC   CONTROL  EQUIPMENT 


is  turned  completely  counterclockwise.  Therefore  the  full  voltage  of  the  lower  transformer 
secondary  is  connected  from  the  cathode  to  the  grid  circuit.  During  every  other  half 
cycle  the  grid  circuit  becomes  positive.  This  charges  the  grid  circuit  capacitor  by  grid 
rectification  in  the  direction  indicated.  The  grid  circuit  is  positive  during  the  half  cycle 


FIG.  3.     General-purpose  Photoelectric  Relays  (Courtesy  General  Electric  Co.) 

in  which  the  plate  is  negative.  Thus  no  plate  current  flows.  When  the  plate  circuit  is 
positive,  the  polarity  of  point  A  is  negative;  also  the  charge  on  the  capacitor  is  in  a  direc- 
tion to  make  the  grid  even  more  negative,  thus  preventing  the  relay  from  being  energized. 
Now,  if  light  is  applied  to  the  phototube,  current  flows  through  the  phototube  and  capacitor 
in  the  direction  of  the  arrow  during  the  half  cycle  in  which  the  grid  is  negative.  This 
effectively  discharges  the  capacitor  in  proportion  to  the  amount  of  light  on  the  phototube. 
If  sufficient  light  is  applied  to  the  photo- 
tube, the  capacitor  charge  due  to  the 
phototube  will  be  greater  than  the  charge 
due  to  grid  current  and  the  grid  will 
become  positive,  thus  energizing  the 
plate-circuit  relay.  When  the  potenti- 
ometer slider  is  in  the  counterclockwise 
position,  the  capacitor  charges  to  a 
higher  value  than  when  it  is  turned  clock- 
wise; therefore  more  light  is  required  to 
operate  the  relay  in  the  counterclockwise 
position  of  the  slider  than  in  the  clock- 


Sbti'onary  operating  vane 


wise  position. 


FIG.  4. 


Circuit  Diagram  of  Relay  4 
Fig.  3 


FIG.    5.      Oscillator-type    Elevator-leveling    Relays 
Mounted  on  an  Elevator  Car  (Courtesy  General  Elec- 
tric Co.) 

^  The  a-c  operated  type  of  photoelectric  relay  just  described  is  not  suitable  for  applica- 
tions where  the  light  impulses  are  of  extremely  short  duration,  because  an  impulse  may 
occur  during  the  time  when  the  sine  wave  of  anode  potential  is  passing  through  zero  and 
therefore  the  relay  would  not  be  energized.  Impulses  of  light  for  a-c  operated  relays  must 
be  at  least  in  excess  of  1  cycle.  In  order  to  operate  relays  where  the  light  impulses  are  of 


RESISTANCE-WELDER  CONTROLS 


21-23 


extremely  short  duration,  it  is  essential  to  use  a  d-c  power  supply  for  the  anode  power  of 
the  amplifier  tube,  as  well  as  for  the  photoelectric  tube  itself.  The  circuit  diagram  of  a 
high-speed  photoelectric  relay  is  shown  in  Fig.  7,  p.  21-17. 

ELECTROMAGNETICALLY  AND  ELECTROSTATICALLY  OPERATED  OSCIL- 
LATOR-TYPE RELAYS.  The  electro- 
magnetically  operated  oscillator-type  relays 
were  probably  first  used  industrially  in  the 
leveling  of  elevators.  Figure  5  shows  a 
typical  installation  of  five  elevator-leveling 
relays  mounted  on  an  elevator  car.  Fig- 
ure 6  shows  the  connection  diagram  for  this 
unit.  When  there  is  no  magnetic  material 
in  the  oscillator  coil  circuit,  the  tube  current 
is  low  and  the  relay  CR  is  de-energized.  As 
the  car  approaches  a  floor,  however,  an  iron  __ 

channel  enters  the  oscillator-coil  magnetic     •&      0    ,-,.      .,    .  ^    .,,  ,  T-,,      ,     ,  """,. 

.       .,        ,        .„    ,.         ,  mi-    •  FIG.  6.    Circuit  of  Oscillator-type  Elevator-leveling 

circuit  and  oscillation  stops.    This  increases  Relay 

the  oscillator  plate  current  and  energizes 

the  relay  CR  which  in  turn  operates  the  necessary  control  equipment  to  stop  the  car  at 

the  desired  level. 

Figure  7  illustrates  the  use  of  an  oscillator-type  relay  which  is  used  in  a  pyrometer  con- 
troller. An  iron  vane  on  the  instrument  pointer  passes  through  the  oscillator  coil  and 
energizes  a  15-amp  control  relay  when  the  pyrometer  reaches  a  predetermined  tempera- 
ture. The  point  at  which  the  oscillator  stops  oscillation  is  very  sharp  and  gives  a  high 
degree  of  control  accuracy.  Also  since  no  mechanical  forces  are  involved,  such  as  there 
would  be  with  mechanical  contacts,  the  accuracy  of  the  instrument  is  not  impaired. 

Electrostatically  operated  oscillating-type  relays  are  used  as  level  controllers  in  grain 
bins  and  in  tanks  which  store  non-conducting  liquids. 


^Control  polofrspottfer 


•pyfometej  pofnter 

inrrdl  circuit  coils 
•Pin  jacks  for  control  coils 

-Load  relay 
Oscillator  coifs 

Vacuum,  tube 


FIG.  7.     Free-vane  Electronic  Pyrometer  Controller  (Courtesy  The  Bristol  Co.) 


12.  RESISTANCE-WELDER  CONTROLS 

The  first  large-scale  applications  of  electronic  control  industrially  were  made  in  the 
resistance-welding  field.  Electronically  controlled  resistance  welders  produced  results  in 
welding  which  were  previously  impossible.  Thus  a  highly  specialized  branch  of  control 


21-24 


ELECTRONIC   CONTROL  EQUIPMENT 


has  been  developed  to  meet  the  many  requirements  of  resistance  welders.  Resistance 
welders  are  divided  into  three  primary  types:  (1)  spot  or  projection  welders;  (2)  seam  or 
roll-spot  welders;  (3)  upset  or  flash  welders.  Most  electronic  welder-control  equipments 
are  used  on  spot,  projection,  and  seam  welders. 

The  sequence  of  operation  in  making  a  spot  weld  is  as  follows:  (1)  The  welder  electrodes 
are  applied  to  the  metal  during  the  "squeeze"  time.  (2)  Welding  current  is  passed  through 
the  material  for  a  predetermined  "weld"  time.  (3)  The  electrodes  are  held  closed  during 
a  "hold"  time,  while  the  material  hardens.  A  pulsation  spot  weld  is  made  in  the  same 
manner  as  a  spot  weld  except  that  during  the  "weld"  time  the  power  is  intermittently 
applied  with  cooling  intervals  between  the  power  intervals.  Seam  welds  are  overlapping 
or  non-overlapping  welds  made  consecutively  by  intermittent  power  pulses  while  the 
material  is  passing  between  two  welding  wheels. 

Most  welder  controls  consist  basically  of  a  thyratron  or  ignitron  contactor  connected 
in  the  primary  of  a  welding  transformer.  Welding  power  levels  vary  from  a  few  hundred 
volt-amperes  for  small  parts  to  values  in  excess  of  1000  kva  for  heavy  parts  and  structures. 
Because  the  resistance  of  the  material  being  welded  is  very  low,  a  very  high  secondary 
current  (1000  to  100,000  amp)  is  required  to  produce  a  weld.  The  relatively  long  secondary 
"loop"  which  is  used,  in  combination  with  the  low  resistance  of  the  secondary  circuit, 
results  in  a  low  power-factor  load. 

From  a  power-system  standpoint,  a  single-phase  load  with  such  high  peak-power 
demands  of  very  short  duration  is  sometimes  a  disadvantage.  Energy-storage-type  welder 
controls  store  energy  either  in  capacitors  or  in  an  inductance  during  the  non-welding  time 
and  "dump"  the  stored  energy  into  the  material  to  produce  a  weld.  These  controls  have 
three  advantages  over  the  conventional  single-phase  controls,  particularly  when  used  in 
welding  aluminum  or  other  high-kva-demand  materials:  (1)  The  control  operates  from  a 
three-phase  power  supply.  (2)  The  peak-power  demand  is  low.  (3)  The  power  factor  is 
higher  than  for  most  single-phase  controls.  Energy-storage  type  of  controls  are  more 
expensive  and  less  flexible  than  single-phase  controls,  and  their  general  acceptance  is 
thereby  limited. 

The  simplest  form  of  electronic  control  for  resistance  welders  is  the  single-phase  thyra- 
tron or  ignitron  contactor  described  previously.  This  provides  high-speed  operation  and 
low  maintenance,  but  it  is  simply  an  on-off  power  control.  Current  starts  to  now  when 
the  initiating  switch  is  closed,  and  it  stops  flowing  at  the  first  current  zero  after  the  initi- 
ating switch  is  opened.  The  energy  in  watt-seconds  which  is  delivered  to  a  weld  =  PUT. 
The  resistance  R  is  dependent  on  the  material  being  welded.  For  a  given  resistance  the 
welding  current  I  is  directly  proportional  to  the  effective  voltage.  With  an  on-off  type 
control  the  current  can  be  changed  only  by  adjustment  of  transformer  taps.  Often  manual 
or  mechanical  timers  are  used  with  ignitron  contactors  to  control  the  time  T. 

In  some  cases  the  approximate  adjustment  of  current  magnitude  provided  by  tap-chang- 
ing combinations  on  the  welding  transformer  are  sufficient.  However,  certain  metals  and 
alloys  which  must  be  welded  rapidly  (within  a  narrow  temperature  range)  require  more 
accurate  current  settings.  Phase-shift  heat-control  equipment  can  be  added  to  ignitron 
contactors  to  provide  these  accurate  settings.  Figure  8  shows  oscillographs  of  welding 
currents  under  three  conditions  of  heat  control. 


Primary 
current 


^ 


A 


V 


V 


Secondary        |/\ 
cur  re  at    """— f — *" 


A 


A 


^A 


V 


V 


Line 
voltage 


A.  A 


V  V 


V  V 


V   V 


«  Low  heat  Moderate  heat  I  Maximum  heat 

FIG.  8.     Oscillograms  of  Current  in  Welder  when  Using  Phase-shift  Heat  Control 

Automatic  weld  timers  are  often  used  with  electronic  contactors  to  coordinate  the 
various  mechanical  operations  as  well  as  to  time  the  welding  power  impulse.  When 
welding  thin-gage  materials,  the  required  time  of  current  flow  may  be  as  short  as  2  cycles. 
A  variation  of  1  cycle  would  mean  a  variation  in  the  heat  input  of  50  per  cent  and  could 


RESISTANCE-WELDER  CONTROLS 


21-25 


FIG.  9.    Oscillogram  Made  When  Current  is  Started  at 
Zero  Point  on  Voltage  Wave 


give  a  faulty  weld.  Therefore  precision-type  timers  are  required.  On  the  other  hand,  if 
the  welding  time  required  were  30  cycles  a  precision  timer  would  not  be  so  important 

rJlnteSr  S68  ^1^^  in^ated  at  a  n°rmal  CUrrent  2ero  a  tra^*t  Current  will  be 
present  which  will  result  m  additional  heat  in  the  weld.  On  short  time  intervals  the 
transient  may  add  considerably  to  the  total  heat.  Variations  in  welds  will  therefore  result 
from  random  firing.  Synchronous  firing  is  often  provided  to  minimize  the  transient  cur- 
rent by  firing  the  power  tubes  at  a  fixed  point  in  the  cycle.  Figure  9  shows  the  transient 
current  which  will  exist  when  current  is  started  at  the  zero  point  on  the  voltage  wave. 
Figure  10  shows  how  the  transient  can 

be  eliminated  by  starting  the  current  at  j*-Start-zero  point  on  voltage  wave 

a  normal  current  zero.  ' 

Figure  11^1  illustrates  a  circuit  which 
will  provide  phase-shift  heat  control  and 
synchronous  firing.  The  thyratron  fir- 
ing circuits  on  both  tubes  are  identical 
except  for  the  instantaneous  polarities 
of  the  transformers  in  the  grid  circuits. 
The  instantaneous  polarities  indicated 
exist  when  the  anode  of  the  left-hand 
thyratron  tube  is  positive.  Figure  1  IB 
shows  the  supply  voltage  E  and  the 
grid  voltage.  The  grid  voltage  consists 
of  two  components:  (1)  EB,  which  is 
the  voltage  of  3T  secondary;  (2)  EP, 
which  is  the  voltage  of  IT  secondary. 
Transformer  IT  is  a  "peaking"  transformer  which  has  only  a  small  amount  of  iron  in 
the  core.  This  results  in  saturation  at  a  particular  point  in  the  wave  and  therefore 
a  peaked  output  voltage,  about  5°  or  10°  wide.  The  phase  position  of  the  voltage 
applied  to  the  peaking  transformer  and  therefore  the  phase  position  of  the  voltage  peak 
is  adjusted  by  the  phase-shifting  rheostat  1R.  The  magnitude  of  Ep  when  added  alge- 
braically to  Eb  is  insufficient  to  fire  the  thyratrons.  Figure  HCt  however,  shows  the  grid 
voltage  when  switch  SW  is  closed.  The  secondary  voltage  Ec  of  2T  is  of  a  polarity  that 
subtracts  from  EB.  Voltage  Ep  is  now  of  a  sufficiently  high  value  to  fire  the  thyratrons. 
During  the  half  cycles  when  the  grids  are  positive,  the  grid  capacitors  are  charged  in  the 
direction  indicated.  This  provides  a  small  negative  grid  bias  to  prevent  misfiring  at  the 
beginning  of  a  cycle  when  the  a-c  grid  voltage  is  going  through  zero. 

A  complete  welder  control  can  be  made  up  of  a  number  of  individual  components,  or 
complete  control  units  can  be  obtained  which  include  some  or  all  of  the  above  features. 
Figure  12  is  an  elementary  diagram  of  a  spot-welder  control  that  provides  precise  timing 
of  the  weld  impulse,  as  well  as  synchronous  firing.  It  does  not  provide  phase-shift  heat 
control.  The  control  essentially  is  made  up  of  a  d-c  control  rectifier  (not  shown),  a 

"keying"  tube  to  insure  starting   the 

j<rStart-m!niroum  transFent  point  power  flow  at  the  desired  point  in  the 

1  on  voltage  wave  voltage  cycle,  a  timing  circuit  which  is 

1  /\  /\  /\  also  a  leading  firing  circuit,  a  trailing 

firing-control  circuit,  and  an  ignitron 
power  circuit.  The  keying  tube  is  nor- 
mally held  non-conducting  by  the  nega- 
tive bias  across  resistance  2.5.  It  is  fired 
at  a  particular  point  in  the  wave  as  de- 
termined by  the  adjustment  of  resist- 
ance 4R  in  series  with  the  primary  of 
the  peaking  transformer.  When  the 
initiating  switch  SI  is  in  the  1  position, 
the  grid  of  tube  2  is  negative.  After 
the  initiating  switch  is  closed  in  the  2 
position,  and  at  the  proper  time  in  the 
next  cycle,  the  grid  of  tube  2  is  made  sufficiently  positive  so  that  it  conducts.  This  in  turn 
fires  the  ignitron  power  tube.  As  capacitor  2C  charges  (the  time  being  dependent  on  the 
setting  of  resistance  6R) ,  the  grid  of  tube  2  becomes  less  positive.  At  a  given  time,  tube  2 
will  again  have  a  negative  grid  voltage  and  therefore  will  not  fire  the  ignitron.  On  an  in- 
ductive load  (a  welder  load),  the  current  in  the  left-hand  ignitron  tube  will  still  be  flowing 
when  the  voltage  applied  to  the  right-hand  ignitron  tube  becomes  positive.  This  means 
that,  as  the  current  in  the  left-hand  ignitron  goes  to  zerp,  the  feedback  transformer  will 
put  a  positive  voltage  on  the  grid  of  tube  3,  thus  making  the  right-hand  ignitron  tube 


FIG.   10.     Oscillogram  Made  When  Current  Flow  is 

Started  at  the  Normal  Current  Zero  by  Synchronous 

Firing 


21-26 


ELECTEONIC  CONTKOL  EQUIPMENT 


Secondary*?        -  EB  + 

3T  Secondary 


A-c  supply 


ST  7   a  2T  ^  Prlmaty 

Primary]         /Prlmaryf 


(A) 


IT 
Secondary 


3T  Secondary 


Vr 
Wio) 


S  open  ^closed 

FIG.  11.     Phase-shift  Heat  Control  and  Synchronous  Firing  is  Provided  by  the  Circuit 


FIG.  12.    Circuit  of  Welder  Control  Having  Precise  Timing  and  Synchronous  Firing,  but  without  Heat 

Control 


D-C  MOTOR  CONTKOL 


21-27 


conduct  for  a  half  cycle  For  each  half  cycle  which  the  left-hand  ignitron  tube  fires, 
there  will  be  duplicate  firing  on  the  right-hand  ignitron  tube.  This  circuit  provides  a 
means  of  controlling  two  tubes  with  different  cathode  potentials  from  a  single  timing 
source.  Furthermore,  it  eliminates  the  possibility  of  obtaining  an  odd  number  of  half 
cycles  of  power,  which  would  cause  saturation  of  the  welding 
transformer.  This  circuit,  however,  cannot  be  used  for  heat 
control. 

Figure  13  shows  a  fully  electronic  welder-control  panel. 
The  top  section  includes  an  electronic  sequence  timer  for 
timing  the  various  parts  of  the  machine  cycle.  The  center 
section  includes  a  synchronous  precision  weld  timer  and  phase- 
shift  heat  control.  The  bottom  section  is  an  ignitron  contactor. 

13.  D-C  MOTOR  CONTROL 

Electronic  adjustable-speed  drives  for  d-c  motors  have  been 
applied  in  many  industries.  Typical  of  these  applications  are : 

(1)  drives  for  several  types  of  machine  tools,  including  lathes, 
grinders,    and  milling  machines;   (2)   paper-machine   drives; 
(3)  rubber-calender  drives;  (4)  conveyor  drives;  (5)  textile- 
range  drives;  (6)  steel-mill  auxiliary  drives;  (7)  testing-equip- 
ment drives. 

In  many  applications  the  entire  motor  power  is  supplied 
through  electronic  rectifiers  of  the  controlled  type.  The  thy- 
ratrons  are  then  usually  controlled  from  vacuum-tube  control 
circuits.  Most  applications  involving  electronic  power  sup- 
plies have  been  made  in  the  low-  and  medium-horsepower 
field,  between  !/4  and  30  hp.  Although  some  applications  of 
electronic  power  supplies  have  been  made  as  high  as  300  hp, 
the  more  conventional  power  converter  in  the  large  size  is  a 
motor-generator  set.  In  some  motor-generator-set  applications, 
thyratron  rectifiers  are  used  to  control  the  field  of  the  genera- 
tor or  the  motor  or  both,  as  necessary.  High-vacuum  tubes 
are  then  generally  used  to  control  the  thyratrons.  In  other 
applications  electronic  control  is  limited  to  the  use  of  vacuum 
tubes  for  controlling  the  low-power  fields  of  pilot  exciters,  such 
as  the  General  Electric  amplidyne,  which  in  turn  control  the 
generator  or  motor-field  power. 

Figure  14  is  a  typical  electronic  control  circuit  which  pro- 
vides adjustment  of  the  armature  voltage  of  a  d-c  motor  by 
thyratron  phase-shift  control  to  obtain  speed  control.  Figure  15  shows  pictorially  the 
essential  components  of  a  complete  adjustable-speed  drive.  This  drive  provides  the  fol- 
lowing features:  (1)  adjustable  motor  speed  over  a  wide  range  from  an  a-c  power  supply; 

(2)  accurate  speed  regulation;  (3)  current-limit  acceleration  to  a  preset  speed;  (4)  com- 
pensation for  the  internal  motor  IR  drop,  regardless  of  motor  load;  (5)  normal  motor 
torque  at  the  instant  of  closing  the  motor-armature  contactor. 

The  motor  field  is  supplied  by  a  non-controlled  rectifier  (not  shown).  D-c  control 
power  is  obtained  from  a  small  control-power  rectifier  (not  shown).  The  a-c  winding  of 
the  saturable  reactor  SR,  the  d-c  winding  of  which  is  shown,  is  connected  in  the 
thyratron  phase-shifting  bridge  to  adjust  the  rectifier  output  voltage  in  proportion  to 
the  amount  of  current  in  the  d-c  winding.  Tubes  1  and  2  are  voltage-regulating  glow 
tubes.  The  speed-control  potentiometer  IP  which  is  connected  across  the  lower  glow 
tube  (tube  2)  provides  a  reference  voltage  against  which  the  motor-armature  voltage  is 
compared. 

Conventional  glow-tube  voltages  are  75,  105,  and  150  volts.  The  d-c  motor  voltages 
are  commonly  230  volts.  Therefore  a  voltage  divider  is  connected  between  lines  24  and  26 
so  that  the  voltages  between  lines  22  and  7  will  be  about  equal  to  the  reference  voltage 
when  the  motor-armature  voltage  is  230.  The  voltage  from  lines  22  to  7  includes  an  IR 
compensation  voltage  which  will  be  discussed  later.  For  the  moment,  however,  assume 
that  the  IR  compensation  potentiometer,  3P,  is  turned  counterclockwise  so  that  26  and  7 
are  at  the  same  potential.  Any  difference  in  voltage  which  exists  between  the  slider  of 
the  speed  control  and  point  22  is  applied  to  the  grid  of  tube  4.  A  capacitor  is  connected 
from  grid  to  cathode  of  this  tube  to  filter  the  rectifier  output  voltage  before  it  is  applied 
to  the  grid. 


L 


FIG.  13.  Fully  Electronic 
Welder  Control  Having  Pre- 
cise Timing  and  Synchronous 
Firing,  but  without  Heat 
Control  (Courtesy  General 
Electric  Co.) 


21-28 


ELECTRONIC  CONTROL  EQUIPMENT 


If  the  armature  voltage  is  proportionally  lower  than  the  speed-control  slider  voltage 
the  grid  of  tube  4  will  be  negative  with  respect  to  the  cathode  and  therefore  tube  4  will 
conduct  no  plate  current.  This  results  in  a  lower  voltage  drop  through  resistor  2R  and, 
consequently,  a  less  negative  voltage  on  the  grid  of  tube  3,  which  increases  the  plate 
current  of  tube  3. 

The  increased  plate  current  which  flows  through  the  d-c  winding  of  SR  increases  the 
rectifier  output  voltage  until  the  grid  voltage  of  tube  4  differs  from  the  cathode  only  by 
the  voltage  required  to  produce  the  desired  rectifier  output  voltage.  A  small  grid-voltage 
change  will  result  in  a  range  of  output  voltage  from  0  to  maximum.  If  the  speed-control 
slider  is  moved,  therefore,  there  exists  momentarily  a  large  differential  voltage  applied 
to  the  grid.  This  will  turn  the  power  tubes  entirely  on  or  off,  as  the  case  may  be,  until 


lurrentVVW  vwJ 


FIG.  14.     Typical  Control  Circuit  for  Electronic  Motor  Control 


the  armature  voltage  has  reached  the  new  level.  Once  it  has  reached  the  new  level  only  a 
small  difference  between  reference  and  signal  voltage  is  needed  to  control  the  rectifier. 

If  the  field  current  of  a  d-c  motor  is  held  constant  the  no-load  speed  is  proportional  to 
the  applied  voltage.  When  mechanical  load  is  applied  to  the  motor  shaft,  however,  the 
armature  current  is  increased.  This  results  in  a  voltage  drop  in  the  internal  motor-arma- 
ture winding  which  reduces  the  speed  in  proportion  to  the  voltage  drop.  A  small  motor 
may  have  a  drop  of  23  volts  at  full  load.  This  is  10  per  cent  of  the  rated  armature  voltage 
of  a  230-volt  motor.  At  rated  applied  voltage  then  there  will  be  a  speed  regulation  of  10 
per  cent  between  no  load  and  full  load.  At  10  per  cent  of  speed  and  no  load,  however,  the 
applied  voltage  is  only  23  volts.  If  this  voltage  is  held  constant  the  speed  will  be  reduced 
to  5  per  cent  at  50  per  cent  of  load  and  further  reduced  to  0  speed  at  100  per  cent  load.  Thi? 
is  obviously  not  a  satisfactory  method  of  operating,  particularly  at  low  speeds. 

Compensation  for  the  IR  drop  is  obtained  in  the  circuit  Fig.  14  in  the  following  manner. 

One  primary  of  a  current  transformer  which  has  two  electrically  separate  primary 
windings  is  connected  in  each  of  the  two  thyratron  rectifier-tube  anode  circuits.  The 
secondary  of  this  current  transformer  is  connected  to  the  rectifier  tube  6.  A  d-c  voltage 
which  is  proportional  to  the  armature  load  current  therefore  appears  across  potentiometer 
2P1  resistance  11 R,  and  potentiometer  3F.  The  polarity  of  the  voltage  across  potentiom- 
eter 3P  is  shown.  Assume  now  that  the  motor  is  running  at  no  load  at  any  desired  speed 
and  that  the  Z-R-compensation  potentiometer  3P  is  turned  clockwise.  Since  no  armature 
current  is  flowing  there  is  no  voltage  across  potentiometer  3P  and  therefore  the  voltage 
from  22  to  7  is  proportional  only  to  the  armature  voltage.  Now,  if  motor  load  is  applied, 
-  Mne  26  will  become  more  negative  than  line  7.  As  a  result  the  grid  of  tube  4  will  also 
become  more  negative.  This  change  in  grid  voltage  acts  through  the  amplifier  to  apply 


D-C  MOTOR  CONTROL 


21-29 


more  armature  voltage.  The  armature  voltage  is  thus  raised  to  compensate  for  the 
resistance  drop  in  the  motor  due  to  load.  Potentiometer  3P  can  be  adjusted  to  compen- 
sate for  the  resistance  drop  in  different  motors  and  therefore  to  hold  the  speed  constant 
with  load  changes.  Capacitor  2C  is  used  to  filter  the  rectifier  output  voltage. 

The  current  transformer  and  rectifier  circuit  also  provide  a  voltage  signal  to  prevent 
overcurrent  which  would  result  in  exceeding  the  rated  peak  current  of  the  power  tubes, 
the  commutating  limit  of  the  motor,  and  perhaps  the  torque  limit  of  the  driven  load. 


FIG.  15.    Pictorial  Elements  of  Electronically  Controlled  D-c  Motor  Drive  (Courtesy  General  Electric 

Co.) 

Assume  that  the  speed  control  is  set  at  a  high-speed  point  and  the  motor-armature  con- 
tactor M  is  closed.  Tubes  4  and  3  will  now  control  the  thyratron  rectifiers  to  give  full 
voltage  output.  This  voltage  would  result  in  approximately  10  times  normal  armature 
current  at  the  first  instant.  The  voltage  between  points  7  and  11  is  proportional  to  the 
armature  current.  If  the  armature  current  is  below  a  normal  value  (approximately  150 
per  cent  of  full-load  current) ,  the  grid  of  tube  5  will  be  negative  with  respect  to  the  cathode. 
If,  however,  it  exceeds  a  normal  value,  the  grid  of  tube  5  will  become  less  negative  and  the 
plate  current  in  tube  5  will  control  the  output  voltage  of  the  rectifier  (in  the  same  manner 
as  tube  4  does)  to  hold  the  accelerating  current  at  a  predetermined  maximum  value. 

When  the  armature  contactor  M  is  open,  the  rectifier  output  voltage  is  still  controlled 
by  the  speed-control  potentiometer.  If  the  speed-control  slider  were  set  for  a  high-speed 
position  there  would  still  be  an  initial  high  current  impulse  when  the  armature  contactor 
first  closed.  To  prevent  this  condition  a  voltage  divider  consisting  of  resistors  7R,  8R, 
and  9JS  is  connected  across  the  regulated  d-c  voltage.  Also  the  rectifier  output  voltage  is 
connected  to  the  divider  at  point  24.  When  the  armature  circuit  is  open,  the  normally 
closed  interlock  on  M  connects  points  45  to  11.  Now,  as  the  rectifier  voltage  is  increased, 


21-30 


ELECTRONIC   CONTROL  EQUIPMENT 


tube  5  draws  plate  current  which  holds  the  rectifier  output  voltage  at  a  level  which  will 
result  in  approximately  normal  armature  current  and  therefore  normal  motor  torque  at 
the  instant  the  armature  contactor  is  closed. 

Many  other  features  can  be  obtained  by  circuit  variations.  Some  common  variations 
are:  (1)  control  of  the  motor-field  voltage  by  thyratron  rectifiers;  (2)  control  of  motor-  or 
generator-field  current;  (3)  more  accurate  speed  control  by  means  of  a  small  permanent- 


PIG.  16.    Electronic  Motor-  and  Generator-field  Control  for  Rubber  Calender  Drive  (Courtesy  General 

Electric  Co.) 

magnet-type  d-c  generator  mounted  on  the  motor  shaft  and  used  as  a  speed  signal;  (4) 
reversing  by  current-limit  regenerative  means;  (5)  speed  programming;  (6)  timed  accelera- 
tion in  lieu  of  current-limit  acceleration;  (7)  motor-speed  control  in  accordance  with  ex- 
ternal signals. 

Figure  16  shows  a  control  panel  for  a  rubber-mill  calender  drive  which  includes  electronic 
control  of  the  fields  of  a  generator  and  a  125-hp  motor. 

14.  SIDE-REGISTER  POSITIONING  CONTROL 

Servomechanism  is  a  term  often  applied  to  the  general  field  of  follow-up  or  positioning 
Side-register  positioning  control  is  one  of  the  many  examples  of  positioning 


SIDE-EEGISTEE  POSITIONING  CONTROL 


21-31 


control  which  could  be  selected  from  this  broad,  general  field.  Side-register  control  is 
applied  to  steel-mill  "cotters,"  to  papermill  "reels,"  and  to  textile-mill  "beams"  to  obtain  a 
smooth  surface  at  the  edge  of  the  finished-material  roll.  It  is  also  used  on  slitters  for 
insulating  tape,  linoleum,  and  the  like  to  guide  material  going  from  a  reel  into  the  slitting 
knives  so  that  only  a  certain  amount  of  undesired  edge  will  be  trimmed,  with  a  minimum 
of  waste  beyond  the  slitter. 

One  frequent  use  is  an  electronic  side-register  positioning  control  applied  to  a  paper 
slitter.  In  this  application  it  is  desirable  to  hold  the  edge  of  the  material  at  a  given  point 
with  respect  to  the  slitting  knives.  A  photoelectric  scanner,  including  a  light  source,  a 
phototube,  and  a  preamplifier,  is  focused  on  the  edge  of  the  paper.  The  photoelectric 
preamplifier  supplies  a  signal  to  the  electronic  control  panel,  which  in  turn  operates  an 
alignment  motor  to  move  a  large  roll  of  paper,  as  well  as  the  nip  roll,  back  or  forward  in 
order  to  obtain  a  fixed  position  of  the  edge  of  the  paper.  Different  types  of  side-register 
positioning-control  equipment  are  available  to  suit  varying  requirements,  such  as  variation 
in  contrast  between  the  material  and  its  background,  range  of  colors  to  be  used,  alignment- 
motor  horsepower,  accuracy  of  positioning,  and  maximum  correction  rate  required. 

The  circuit  diagram  shown  in  Fig.  17  is  that  of  a  simple  on-off  type  control.  This  con- 
trol with  its  associated  reversible  a-c  motor  and  mechanical  brake  will  meet  the  require- 
ments of  applications  where:  (1)  there  is  a  reasonably  large  light  contrast  between  the 


2CR, 


10H 


FIG.  17.     On-off  Type  Photoelectric  Side-register  Positioning-control  Circuit 

material  to  be  positioned  and  its  background  or  where  direct  light  transmission  can  be 
used;  (2)  the  required  accuracy  is  not  greater  than  plus  or  minus  Vie  in.;  (3)  the  required 
rate  of  correction  is  not  greater  than  5  in.  per  min.  The  correction  motor  in  this  case 
is  a  reversible  a-c  motor  which  is  operated  in  the  forward  or  reverse  direction  by  magnetic 
contactors,  which  are  in  turn  operated  by  relays  ICR  and  2CR.  If  there  is  no  light  shining 
on  the  phototube,  the  grid  of  tube  1  will  be  negative.  If  tube  1  is  not  drawing  current, 
the  cathode  will  go  slightly  negative,  but,  in  doing  so,  since  the  cathode  of  tube  2  is  con- 
nected to  the  cathode  of  tube  1,  tube  2  will  be  turned  on.  The  lack  of  current  in  tube  1 
will  cause  tube  4  to  conduct  and  energize  relay  \CR.  Since  tube  2  is  conducting,  tube  5 
will  be  non-conducting  and  relay  2CR  will  be  de-energized.  If  a  large  amount  of  light  is 
now  applied  on  the  phototube,  the  grid  of  tube  1  will  be  positive  and  tube  2  will  conse- 
quently be  turned  off.  This  will  result  in  the  opposite  operation  of  relays  ICR  and  2CR. 

The  light-intensity  adjustment  can  be  turned  clockwise  to  decrease  the  intensity  that 
will  cause  the  relays  to  operate.  For  a  given  light  intensity,  the  centering  adjustment  will 
make  the  transition  in  the  relays  occur  when  approximately  half  of  the  light  beam  is 
intercepted  or  reflected,  depending  on  the  method  of  light  transmission.  It  is  obvious 
that,  if  there  were  no  dead  zone  between  the  operation  of  the  two  control  relays,  the  equip- 
ment would  be  continually  hunting  from  one  side  to  the  other.  The  dead-zone  adjustment 
provides  an  adjustable  amount  of  distance  over  which  the  material  can  travel  without 
actuating  the  control  relays. 

In  applications  where  continuous  operation  of  the  correcting  motor  is  desirable,  the 
system  just  described  is  not  suitable  because  of  the  maintenance  of  the  mechanical  parts. 
In  these  applications  the  photoelectric  amplifier  can  control  two  half-wave  thyratron  rec- 
tifiers which  are  connected  in  inverse  parallel  to  operate  a  shunt-wound  d-c  motor  in  the 
forward  or  reverse  direction.  For  correcting  motors  of  3/4  hp  or  larger,  vacuum  tubes  are 


21-32 


ELECTRONIC   CONTROL  EQUIPMENT 


generally  used  to  operate  the  field  of  a  pilot  generator  which  will  in  turn  supply  power  to 
operate  the  correcting  motor  in  the  forward  or  reverse  direction.  Where  wide  ranges  of 
light  intensities  and  small  light  differentials  are  to  be  encountered,  a  more  elaborate  con- 
trol equipment  involving  a  rotary-lens-type  scanning  head  is  often  used. 


15.  PROCESS  CONTROLS 

Electronic  control  has  recently  made  its  way  into  the  field  of  process  instrumentation 
and  control,  where  it  is  now  frequently  used  in  measuring,  recording,  and  controlling  such 
process  variables  as  temperature,  pressure,  flow,  pH,  moisture,  and  the  like.  An  electronic 
control  unit  included. as  a  part  of  a  complete  potentiometer  controller  is  shown  semi- 
schematically  in  Fig.  18.  The  battery  supplies  a  standard  reference  voltage  to  the  slide 


SLIDE    WIRE    (E) 


TEMPERATURE  SCALE  (F) 


CONVERTER 


BATTERY  /    * 
THERMOCOUPLE 


A.  C.  LINE 


FIG.  18.     Semischematic  Diagram  of  Continuous-balance  Potentiometer  Controller  (Courtesy  Brown 

Instrument  Co.) 

wire.  The  voltage  produced  by  the  thermocouple  is  compared  with  the  voltage  at  the 
slider.  Since  the  voltage  magnitudes  are  so  small,  however,  it  is  not  suitable  to  apply  this 
voltage  difference  directly  to  a  d-c  amplifier.  Instead,  it  is  connected  to  a  vibrating-type 
inverter  which  is  synchronized  with  the  power-supply  frequency  to  produce  a-c  voltage 
impulses  of  a  given  phase  relation  in  the  amplifier  tubes.  If  a  difference  exists  between  the 
thermocouple  voltage  and  the  voltage  of  the  standard,  this  signal  is  amplified  by  an  a-c 
voltage  amplifier  and  further  by  means  of  vacuum-type  power-amplifier  tubes  to  control 
the  power  applied  to  a  small  reversible-type  a-c  balancing  motor.  The  reversible  balancing 
motor  repositions  the  slider  until  the  thermocouple  voltage  matches  that  of  the  battery. 
Thus,  the  pointer  is  positioned  to  show  the  temperature  which  exists  at  the  thermocouple 
location. 

If  pressure  is  to  be  measured  or  controlled,  the  pressure-sensitive  element  can  be  a 
variable  reactance.  This  can  be  used  in  an  a-c  bridge  circuit,  and  the  mechanical  inverter 
is  not  necessary. 

16.  SYSTEM  STABILIZATION 

In  any  electromechanical  system  there  is  a  time  lag  between  the  application  of  a  correc- 
tion signal  and  the  final  corrected  value;  for  example:  (1)  if  a  given  voltage  is  applied  to  the 
field  of  a  generator,  the  inductance  of  the  field  prevents  the  field  current  and  therefore  the 
generator  output  voltage  from  building  up  instantaneously;-  (2>  if  a  given  voltage  is  applied 
to  the  armature  of  a  d-c  motor  the  mechanical  inertia  prevents  the  motor  speed  from  instan- 
taneously reaching  its  new- value;  (3)  if  a  correction  voltage  is  applied  to  the  correcting 
motor  of  a  servomechanism  positioning  control  not  only  must  the  correcting  motor  accel- 
erate its  own  inertia  plus  the  inertia  of  the  connected  load  but  it  must  also  run  at  rated 
speed  until  the  new  position  is  reached — then  it  must  decelerate. 


SYSTEM  STABILIZATION 


21-33 


Assume  that  a  sensitive  and  instantaneously  operating  electronic  voltage  regulator  is 
applied  to  a  d-c  generator.  If  the  generator  output  voltage  is  lower  than  that  called  for 
by  the  voltage-adjusting  potentiometer,  the  regulator  will  immediately  apply  the  maximum 
field  voltage  to  attempt  to  correct  the  output  voltage.  The  output  voltage,  however,  will 
increase  slowly  because  of  the  field  inductance.  However,  when  the  output  voltage  reaches 
the  desired  value  the  electronic  regula- 
tor immediately  applies  the  correct  field 
voltage  to  hold  the  desired  output  volt- 
age. Such  a  system  may  be  slow  in  re- 
sponse, but  it  is  stable  in  its  operation. 

If  on  the  other  hand  it  is  assumed  that 
the  electronic  regulator  has  internally  an 
inherent  time  lag,  the  operation  of  the 
entire  system  will  be  quite  different.  If 
the  generator  voltage  is  low,  the  regu- 
lator will  again  apply  maximum  field 
voltage.  When  the  generator  voltage 
reaches  the  desired  point,  the  regulator 
will  not  immediately  apply  the  correct 
field  voltage  because  of  the  inherent 
time  lag  in  the  regulator.  The  correct 
field  voltage  will  not  be  established  until 


Method  of  Adding  a  Predominant  Time  Lag 
to  Eeduce  System  Hunting 


the  output  voltage  has  exceeded  the   FIG.  19. 
desired  value.    Now,  however,  the  regu- 
lator receives  a  signal  which  will  result 
in  a  weaker  field  than  normal.    As  a  result,  the  system  may  be  unstable  and  continue  to 
oscillate  above  and  below  the  desired  voltage  level. 

The  frequency  of  oscillation  or  hunting  is  determined  by  the  time  constants  of  a  system. 
If  long  time  constants  are  involved,  the  hunting  will  be  at  a  low  frequency;  if  short  time 
constants  are  involved,  the  hunting  will  be  at  a  high  frequency.  In  order  for  hunting  to 
exist  in  any  system,  two  conditions  must  exist:  (1)  the  total  time  lags  of  the  entire  regulat- 
ing system  must  result  in  the  regulating-voltage  signal  being  fed  back  to  the  regulator 
input  180°  out  of  phase  with  the  normal  sense  of  the  correction  voltage;  (2)  at  the  fre- 
quency at  which  the  total  time  lags 
add  up  to  180°,  the  overall  system 
amplification  must  be  equal  to  or 
greater  than  1. 

Obviously  hunting  is  undesirable  in 
most  systems.  Therefore  means  must 
be  established  to  eliminate  it.  Several 
methods  are  in  common  use. 

As  was  pointed  out  earlier,  a  system 
with  a  single  time  constant  will  not 
hunt.  (Although  this  is  true  in  general 
there  are  single-lag  systems  which  will 
hunt.)  Hunting  in  a  system  having 
two  or  more  independent  time  lags  can 
be  reduced  by  artificially  adding  a  time 
constant  which  is  sufficiently  greater 
than  the  others  that  it  predominates. 
Such  a  system  will  approach  a  single- 
lag  system  in  performance.  Figure  19 
shows  how  a  lag  can  be  added  to  a 
d-c  amplifier  circuit.  The  grid-to- 
plate  capacitor  charges  or  discharges 
through  the  series  grid  resistor  at  a 
rate  dependent  on  the  circuit  con- 
stants. The  action  of  the  grid  there- 
fore lags  behind  the  input  signal. 


FIG.  20.     Method  of  Adding  a  Derivative  or  Rate-of- 
change  Signal  to  Reduce  System  Hunting 


Since  the  overall  amplification  of  the  system  must  be  at  least  1  at  the  hunting  frequency, 
a  simple  way  of  reducing  hunting  is  to  reduce  the  system  amplification.  Conversely,  the 
higher  the  system  amplification  is,  the  more  likely  the  system  is  to  hunt. 

Often  rate  of  change  or  derivative  signals  may  be  added  to  the  input  signal  to  reduce 
hunting.  Figure  20  shows  a  typical  circuit  used  in  a  speed  regulator.  The  steady-state 
grid- voltage  signal  from  the  tachometer-generator  which  is  connected  to  the  motor  shaft 


21-34  ELECTRONIC   CONTROL  EQUIPMENT 

is  determined  by  resistors  2R  and  3R.  During  a  change  in  speed,  however,  the  capacitor  C 
supplies  a  grid  voltage  which  is  proportional  to  the  rate  of  change  of  speed  and  thus  pro- 
vides a  grid  signal  which  is  ahead  in  time  relation  to  the  speed  signal. 

Rigorous  solutions  to  stability  problems  are  often  complex.  Furthermore,  assumptions 
of  system  constants  or  simplifying  assumptions  often  are  proved  to  be  in  error  when  equip- 
ment is  installed.  As  a  result  most  systems  are  stabilized  by  empirical  methods. 

BIBLIOGRAPHY 

Chin,  P.  T.,  and  E.  E.  Moyer,  Principles  of  Grid  Control  for  Thyratrons,  A.I.E.E.  Technical  Paper 
45-62  (January  1945). 

Chin,  P.  T.,  and  E.  E.  Moyer,  A  Graphical  Analysis  of  the  Voltage  and  Current  Wave  Forms  of  Con- 
trolled Rectifier  Circuits,  Electrical  Engineering,  July  1944. 

Chute,  G.  M.,  Electronics  in  Industry,  McGraw-Hill  (1946). 

Chute,  G.  M.,  Fundamentals  of  Industrial  Electronics,  Steel,  April  3-May  22,  1944. 

Chute,  G.  M.,  Resistance  Welding  Control,  McGraw-Hill. 

Cockrell,  W.  D.,  Grid  Control  of  Gas  Filled  Tubes,  Electronics,  June  1944. 

CockreU,  W.  D.,  Industrial  Electronic  Control,  McGraw-Hill  (1944). 

Cooper,  B.,  Better  Welds  through  Regulated  Welding  Current,  Welding  Journal,  January  1944. 

Dalton,  B.  J.,  Electronic  Motor  Control,  Proceedings  of  National  Electronics  Conference,  1944. 

Dalton,  B.  JM  Electronic  Motor  Control,  Gen.  Elec.  Rev,,  May  1945. 

DeBlieux,  E.  V.,  Characteristics,  Design  and  Application  of  Rectifier  Transformers,  Gen.  Elec.  Rev., 
October,  November,  December  1937. 

Hall,  James  H.,  Transformer  Calculations  for  Selenium  Rectifier  Applications,  Electrical  Manufacturing, 
February  1946. 

Hazen,  H.  L.,  Theory  of  Servo-Mechanisms,  Journal  of  the  Franklin  Institute,  September  1943. 

Kloeffler,  R.  G.,  Principle*  of  Electronics,  Wiley  (1942). 

Leigh,  H.  HM  and  H.  L.  Palmer,  Inverter  Action  on  Reversing  of  Thyratron  Motor  Control,  Electrical 
Engineering,  April  1944. 

M.  I.  T.  Staff,  Applied  Electronics,  McGraw-Hill. 

Moyer,  E.  E.,  Electronic  Control  of  D-c  Motors,  Electronics,  May,  June,  July,  September,  October  1943. 

Palmer,  H.  L.,  M.  E.  Bivens,  and  S.  A.  Clark,  Electronic  Welding  Control,  Electronics,  August,  Septem- 
ber, October  1943. 

Reich,  H.  J.,  Theory  and  Applications  of  Electron  Tubes,  McGraw-Hill  (1939). 

Smith,  E.  S.,  Automatic  Control  Engineering,  McGraw-Hill  (1944). 


SECTION  22 
AIDS  TO  NAVIGATION 


RADIO  AIDS  TO  AIR  NAVIGATION 

ART.  BY  HENRY  I.  MUTZ  PAGE 

1.  Introduction 04 

2.  Terminology  and  Definitions 05 

3.  Radio  Aids  in  the  Federal  Airways  Sys- 

tem Today 06 

4.  Facilities  in  the  New  Federal  Airways 

System 13 

5.  Proposed  New  Landing  Systems 26 

6.  Proposed  New  Short-range  Navigation 

Systems 28 

7.  Proposed   New   Long-range   Navigation 

^      Systems 31 


AST.  PAGE 

8.  Miscellaneous  Radio  Aids 31 


RADIO  AIDS  TO  MARINE  NAVIGATION 

BY  M.  K.  GOLDSTEIN  : 

'  9.  Established  Navigational  Aids 36 

10.  Recently  Introduced  Navigational  Aids.     44 

11.  Contemplated    and    Proposed    Naviga- 

tional Aids 63 

12.  Determination  of   Optimum    Transmis- 

sion Parameters  for  Some  Long-range 
Radio  Navigation  Systems 57 


22-01 


FIG.  1.     The  Present  35,000  Miles  of  Federal  and  Civil  Airways.     The  airway 


22-02 


CIVIL  AIRWAYS.  AIRWAY  TRAFFIC  CONTROL 

AREAS.  AND  FLIGHT  ADVISORY  AREAS  OF 

THE  UNITED  STATES  AND  CANADA 

DATE:  JULY  1.  IMS 

CIVIL  AERONAUTICS  ADMINISTRATION 

DEPARTMENT  OP  COMMERCE 

WASHINGTON.  CX  C. 


rUfflJT  LEVELS 

max  AND  «o  aw.  «*»ws 

MTMUNO  000  THOUSAND  FOOT  U«t* 
WESTBOUND  EVEN  THOUSWD  FOOT  LEVBS 

AM  MR  ANO  BLUE  OTl.  MKWAYS 
NORTHBOUND  000  THOUSWD  fOOT  LEVCU 
KMIHiOUNO-iyO«  TMOUSMIO  TOOT  lEVtti 


traffic  control  boundaries  and  centers  are  also  shown.     (Courtesy  CAA.) 


22-03 


AIDS  TO  NAVIGATION 
RADIO  AIDS  TO  AIR  NAVIGATION 

By  Henry  I.  Metz 

1.  INTRODUCTION 

Thirty-five  thousand  miles  of  federal  airways  exist  today  in  the  airspace  over  the  United 
States.  This  mileage,  shown  in  Fig.  1,  is  constantly  and  very  rapidly  increasing.  Informa- 
tion about  the  condition  of  the  airway,  weather  ahead,  and  other  traffic  is  available  con- 
stantly to  the  pilot  by  means  of 
automatic  radio  aids  to  naviga- 
tion and  two-way  radio  voice 
communications. 

The  establishment,  operation, 
and  maintenance  of  the  airways 
are  among  the  functions  of  the 
Civil  Aeronautics  Administration 
(CAA)  under  the  U.  S.  Depart- 
ment of  Commerce. 

When  the  federal  airways  pro- 
gram was  started  in  1926,  the  de- 
velopment of  a  reliable  radio 
communication  and  guidance  sys- 
tem was  undertaken.  Basically, 
tracks  were  established  in  the  air- 
Fro.  2 (a) .  View  of  Aircraft  Automatic  Direction  Finder  Equip- 


ment Snowing  the  Loop  Antenna  (Courtesy  Bendix  Aviation 
Corporation) 


space  by  overlapping  keyed  radio 
patterns,  the  points  of  overlap 
being  interpreted  aurally  by  the 
pilot,  who  wears  a  pair  of  headphones.  Although  originally  produced  by  a  pair  of  crossed 
loop  antennas,  the  tracks  are  today  made  by  an  Adcock  array  of  vertical  radiators  which 
give  greater  night-time  stability.  The  transmitting  station  is  called  a  "radio  range  station." 
It  produces  four  tracks  called  "courses,"  all  of 
wnich  emanate  radially  in  predetermined  fixed 
directions.  There  are  399  range  stations  now  in 
operation.  The  coordinated  alignment  of  a  series 
of  range  courses  constitutes  an  airway. 

A  system  of  markers  has  been  added  to  the 
range  courses.  "These  are  vertically  directed 
signals  at  75  Me,  received  on  a  special  receiver 
in  the  airplane  and  connected  to  a  signal  light 
on  the  instrument  panel.  The  light  operates 
only  when  the  airplane  is  over  the  marker 
station. 

An  automatic  direction  finder  (ADF),  as 
shown  in  Fig.  2,  is  carried  today  by  most  com- 
mercial airplanes  and  is  required  (in  some  form) 
by  CAA  on  scheduled  airliners.  Its  pointer  indi- 
cates the  direction  to  the  station  tuned  in.  It 
may  be  used  as  an  aid  in  flying  the  range  course 
or  to  determine  position  by  taking  bearings  on 
two  or  more  stations. 

Countries  other  than  United  States,  Canada,  ing 
and  Australia  have  based  their  traffic  opera- 
tions on  ground  station  direction  finding  (DF).  DF  stations  are  available  in  these 
countries  to  give  bearings  to  the  aircraft  calling,  just  as  is  now  done  in  locating  position 
on  ships  at  sea. 

22-04 


FIG.  2(&).  View  of  Aircraft  Automatic  Di- 
rection Finder  Equipment  Showing  the  Bear- 
Indicator  (Courtesy  Bendix  Aviation 
Corporation) 


TERMINOLOGY  AND  DEFINITIONS  22-05 

Airborne  direction  finders  have  also  been  used  extensively  in  other  countries  in  coopera- 
tion with  high-powered,  non-directional,  transmitting  stations  on  the  ground.  A  system 
called  "radio-phare"  has  been  employed  in  which  three  spaced  ground  stations  transmit 
in  time  sequence  on  the  same  frequency.  Without  readjustment  of  the  airborne  ADF 
receiver,  three  bearings  can  be  quickly  obtained  from  the  radio-phare.  The  German  33- 
Mc  "Lorenz"  instrument  landing  system  (localizer,  glide  path,  and  markers),  in  particular 
the  localizer,  was  in  quite  general  use  before  the  war.  Its  localizer,  using  interlocked  aural 
dots  and  dashes  (similar  to  the  U.  S.  Aural  AN  Radio  Range)  to  differentiate  left  from  right 
in  approaching  the  runway,  was  used  for  distances  up  to  100  miles  in  cases  where  the 
fixed  course  alignment  agreed  with  the  desired  flight  direction. 

The  CAA  is  engaged  in  a  program  of  converting  all  the  federal  airways  aids  to  static-free 
VHF,  except  that  a  few  strategically  located  low-frequency  range  stations  will  be  retained 
(with  increased  power)  to  serve  for  long  cross-country  nights.  In  addition  to  converting 
to  VHF,  all  future  ranges  will  be  of  the  "visual"  type  (i.e.,  using  pointer  instruments 
instead  of  headphones),  and  will  have  omnidirectional  courses  (i.e.,  instead  of  only  four 
range  courses,  the  pilot  may  select  any  radial  track,  or  course,  from  the  station)  to  take 
care  of  increased  numbers  of  airplanes  and  airports.  Instrument  landing  systems,  de- 
veloped previously  and  now  installed  at  110  airports,  will  be  used  at  once  to  relieve  airport 
traffic  congestion  in  bad  weather.  Radar  will  be  used  soon  as  a  traffic  surveillance  device 
and,  perhaps  later,  as  a  direct  means  of  controlling  traffic.  Pulse  techniques  will  probably 
be  applied  immediately  in  new  meter-type  distance-measuring  equipment  (DME)  and 
later  in  anticollision  devices.  Long-distance  nights  across  water  will  be  guided  by  rela- 
tively low-frequency  aids  such  as  omnidirectional,  Loran,  Sonne,  or  other  systems. 

Automatic  flight,  controlled  by  radio  signals  transmitted  from  the  ground,  has  already 
been  demonstrated  to  be  more  accurate  than  human-pilot-controlled  flight.  It  will  un- 
doubtedly be  used  by  all  commercial  aircraft  in  conjunction  with  automatic  computers, 
to  permit  controlled,  track-type  flying  in  any  direction  regardless  of  station  position. 

2.  TERMINOLOGY  AND  DEFINITIONS 

Certain  terms  associated  with  radio  aids  to  air  navigation  in  engineering  discussions 
are  listed  below  with  their  definitions: 

Radio  Range.  Any  CW  radio  station  whose  radiation  inherently  produces  directional 
courses,  or  tracks,  fixed  in  their  relation  to  the  earth's  surface  and  independent  of  aircraft 
heading.  A  radar  range  is  similar  but  uses  pulse  radiation. 

Marker.  Any  station  having  limited  or  directed  radiation  used  to  give  an  aircraft  its 
position  along  a  range  course. 

Heading.  Direction  (azimuthal  and  clockwise  from  north)  in  which  aircraft  is  point- 
ing. It  agrees  with  direction  of  flight  if  there  is  no  crosswind. 

Bearing.  Azimuthal  angle  (from  north,  clockwise)  of  line  between  fixed  ground  station 
and  airplane,  or  vice  versa.  It  is  generally  necessary  to  state  where  the  bearing  is  from 
and  to,  in  order  to  avoid  ambiguity. 

Track.  Actual  direction  of  motion  of  aircraft  with  respect  to  earth's  surface,  expressed 
in  degrees  of  azimuth  (from  north) . 

Beacon,  Non-directional.  A  radio  transmitting  station  whose  radiation  is  essentially 
uniform  in  all  directions  or  which  does  not  use  directional  radiation  characteristics  to  con- 
vey intelligence. 

Beacon,  Omnidirectional.  A  beacon  whose  directional  or  other  radiation  characteristics 
cause  it  to  give  information  equally  in  all  directions. 

Course  Sharpness.  The  relation  between  angular  displacement  from  course  and  deflec- 
tion of  the  pointer  of  the  indicating  instrument,  usually  expressed  in  angular  degrees  of 
displacement  required  to  give  full-scale-left  to  full-scale-right  pointer  movement.  This 
sharpness  is  generally  a  function  of  ground-station  pattern  sharpness  and  receiver  gain 
setting.  Localizers  are  generally  used  with  4°  to  5°  sharpness. 

Pattern  Sharpness.  The  difference  in  pattern  amplitudes  at  a  given  angular  displace- 
ment from  the  equal  or  on-course  line,  usually  expressed  in  decibels  per  1.5°.  Standard 
figure-of-eight  patterns  (LF  aural  ranges)  give  0.45  db  per  1.5°.  The  CAA  localizer 
patterns  give  about  5  db  sharpness. 

Clearance.  The  db  difference  in  patterns  producing  a  course,  at  angles  other  than 
those  containing  the  on-course.  High  clearance  is  desirable  so  that  the  indicator  will 
remain  fully  deflected  everywhere  except  at  the  course. 

Multiples.  Extra  or  abnormal  courses  resulting  from  zero  or  negative  pattern  clearance 
or  from  severe  reflection  of  signal  from  buildings,  trees,  etc. 


22-06  AIDS  TO  NAVIGATION 

Bends.  Angular  deviation  or  distortion  of  the  on-course  signal  from  a  true,  straight 
radial  line  from  the  station.  Generally,  bends  are  produced  by  reflection  of  the  signal 
from  buildings  or  wires  near  the  transmitting  station.  Bend  magnitude  is  proportional 
to  the  ratio  of  reflected  signal  to  direct  signal  amplitude  and  inversely  proportional  to 
pattern  sharpness.  Apparent  bends  can  be  caused  by  poor  receiver  AVC  action,  allowing 
inferior  circuit  components  to  unbalance  as  signal  strength  changes  with  distance. 

Scalloping.  The  irregular  or  wavy  shape  of  an  antenna  field  pattern  caused  by  reflec- 
tion of  the  signal  from  ground  objects.  Scalloping  is  evidenced  by  the  periodic  hesitation 
in  the  course  indicator  movement  as  the  airplane  is  flown  across  the  course.  Scalloping 
indicates  bends  when  it  occurs  near  the  on-course  line.  When  scalloping  is  severe,  multiple 
courses  are  produced. 

Wiggles.  Rapid,  random,  and  erratic  movement  of  the  course-indicating  pointer, 
generally  caused  by  combined  signal  from  several  reflecting  objects,  especially  trees.  Poor 
electrical  connections  and  noise  also  cause  wiggles.  Wiggles  generally  do  not  alter  the 
average  course  line  and  can  therefore  be  filtered  out  of  the  indicator. 

Pushing  (or  pulling).  Displacement  of  indicated  course  with  heading  of  airplane. 
Term  is  derived  from  observation  in  cross-course  flights  that  the  course  was  apparently 
"pushed"  ahead  of  the  airplane.  Pushing  is  caused  by  the  radiation  of  impure  polarization 
(vertical  in  a  horizontally  polarized  system,  and  vice  versa) .  Attitude  effect,  in  which  the 
indicator  shifts  with  airplane  roll,  or  pitch,  is  similar  in  cause  to  pushing. 

Distance  Range.  Distance  in  miles  from  the  station  at  which  useful  signal  is  lost,  or 
where  course  sharpness  decreases  (loss  of  AVC).  For  VHF  and  above,  the  line  of  sight 
range  generally  prevails. 

3.  RADIO  AIDS  IN  THE  FEDERAL  AIRWAYS  SYSTEM  TODAY 

The  present  system  consists  of  the  following  facilities  or  services : 

A.  Radio  ranges,  LF  (four-course  aural  with  simultaneous  voice) . 

B.  Radio  ranges  (visual  two-course,  VHF). 

C.  Radio  markers  (75  Me). 

D.  Automatic  direction  finder  (ADF)  receivers. 

E.  Communications  (HF  and  VHF). 

F.  Air  traffic  control  and  weather  reporting. 
Each  of  these  is  explained  separately  below. 

RADIO  RANGES,  LF  (four-course  aural  with  simultaneous  voice).  The  radio  range 
of  today  is  a  400-watt,  highly  developed,  four-course  facility  on  which  practically  all  civil 
air  navigation  is  based.  It  is  notr  a  perfect  device  but  is  simple  and  effective  for  distance 
ranges  up  to  several  hundred  miles.  Its  irregularities  are  so  well  known  that  many  of 
them  appear  to  the  trained  pilot  as  an  asset.  The  change  in  conductivity  of  the  earth 
along  the  course,  in  some  places,  usually  in  mountainous  terrain,  causes  bends  and  multiple 
courses  to  appear.  The  location  of  these  multiples  is  known  and  is  plotted  on  charts. 
Night  effect,  which  is  variable,  has  been  minimized  by  the  replacement  of  loop-type 
transmitting  stations  with  Adcock  vertical  tower  systems.  Two  highly  important  objec- 
tions to  the  range  are  (1)  interference  by  atmospheric  and  precipitation  static,  and  (2) 
interference  from  other  stations  in  the  allotted  frequency  spectrum. 

The  present  four-course  Adcock  antenna  system  consists  of  five  steel  towers  about  130 
ft  high.  Four  towers  are  placed  on  the  corners  of  a  square;  diagonally  opposite  towers 
constitute  a  pair  and  are  about  600  ft  apart.  The  fifth  tower  is  at  the  center.  The  pairs 
are  connected,  but  with  reversed  phase,  so  that  they,  respectively,  radiate  figure-of-eight 
horizontal  patterns.  The  two  pairs  are  connected  to  a  crystal-controlled  transmitter 
through  the  equivalent  of  a  single-pole,  double-throw  relay  called  a  "link  circuit  relay." 
The  relay  is  operated  by  a  motor-driven  keyer  unit  so  that  one  pair  of  towers  gives  a 
series  of  dot-dash  (A  in  code)  characters.  The  back  contact  of  the  relay  causes  the  recip- 
rocal character,  dash-dot  (N  in  code)  to  be  keyed  into  the  other  pair  of  towers.  The 
schematic  diagram  is  shown  in  Fig.  3. 

The  two  figure-of-eight  patterns  thus  contain  reciprocal  or  interlocking  characters.  To 
an  observer  with  a  receiver  in  space  the  character  heard  would  depend  on  position  around 
the  station.  In  a  position  where  the  patterns  overlap,  that  is,  where  the  A  and  N  patterns 
are  of  equal  amplitude,  no  character  would  be  observable  because  they  are  interlocked 
and  the  signal  is  a  steady  tone.  The  patterns  and  courses  are  shown  in  Fig.  4. 

Actually,  the  radiation  from  the  two  pairs  of  towers  is  unmodulated  carrier  energy  and 
therefore  inaudible.  The  center  (fifth)  tower  is  connected  to  a  second  crystal-controlled 
transmitter,  but  its  carrier  frequency  is  1020  cycles  below  that  used  for  the  diagonal  towers. 
Consequently,  in  combination,  the  receiver  produces  a  1020  cycle-output  resulting  from 


RADIO  AIDS  IN  FEDERAL  AIRWAYS  SYSTEM  TODAY     22-07 


East 


West  Center          North 


Goniometer 


MIc 


:  Wire  line 
I  &  Strowger 
i  control 


Dual 
transmitter 


FIG.  3.     Schematic  Diagram  of  Four-course  Aural  LF  Radio  Range  Showing  Simultaneous  Voice 

N 


Fan  marker 
keying. 


Fan  marker 

keying etc. 


'  A"  quadrant 

rZ"  marker  (75  Mcs) 


Fan  marker 

75  Mcs,  3,000  cps 

keying etc. 


Fan  marker 

keying etc* 

FIG.  4.     Patterns  and  Courses  of  LF  Aural  Radio  Range  Showing  Flexibility  of  Course  Displacement 

and  Position  of  75-Me  Markers 


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AIDS  TO  NAVIGATION 


the  beat  between  the  two  carriers.  In  effect,  the  center  tower  radiates  the  carrier  and  the 
receiver  is  tuned  to  it.  The  signal  from  the  side  towers  constitutes  sideband  energy.  The 
power  of  the  side  tower  transmitter  is  adjusted  to  give,  30  per  cent  of  that  of  the  center 
tower  carrier  signal  along  the  course.  The  remaining  70  per  cent  carrier  is  used  for  trans- 
mission of  voice  modulation.  The  airborne  receiver  generally  contains  a  combination 
band-pass,  band-rejection  filter  so  that  the  pilot  may  select  voice  or  range  signals  without 
interaction. 

The  course  of  the  aural  AN  range  is  normally  about  3°  wide.  The  width  depends  upon 
headphone  level  and  the  pilot's  hearing  ability.  The  change  from  on-course  to  full  off- 
course  is  gradual;  that  is,  a  "twilight"  zone  exists  to  either  side  of  the  course  which  permits 
the  pilot  to  estimate  (very  approximately)  his  nearness  to  the  course. 

Normally,  four  courses  are  produced  at  right  angles  to  each  other.  All  four  courses 
may  be  rotated  equally,  clockwise  or  reverse,  by  turning  a  goniometer  through  which  the 
outside  antennas  are  connected.  Or,  if  other  than  right-angle  courses  are  required,  as 
shown  in  Fig.  4,  the  relative  phase  of  the  current  in  opposite  towers  is  varied  by  means  of 
artificial  lines.  The  total  line  length  between  towers  is  maintained  at  an  optimum  value 
for  maximum  phase  and  current  stability.  Under  optimum  line  conditions,  the  detuning 
of  rain,  ice,  and  snow  on  one  antenna  reacts  to  produce  an  equal  effect  on  the  other  an- 
tenna, giving  greatest  stability  of  courses. 

RADIO  RANGES  (visual  two-course,  VHF).  The  development  of  a  two-course  VHF 
radio  range  was  started  in  1928  to  overcome  the  static,  congestion,  and  dangerous  quadrant 
ambiguity  problems  existing  on  the  conventional  four-course,  low-frequency,  aural  range. 
Now  several  airways  are  operating  (temporarily  on  the  110-Mc  frequency  band)  with  this 
type  of  facility,  pending  conversion  of  the  entire  federal  airways  to  the  VHF  omnidirec- 
tional system. 

The  complete  designation  of  the  two-course  range  is:  "VHF  two-course  visual  radio 
range  with  quadrant  identification."  This  designation  signifies  particularly  that  the 
quadrant  ambiguity  of  the  four-course  type  has  been  eliminated.  Actually,  the  quadrant 
identification  comes  through  the  superposing  of  two  aural  courses  on  the  same  visual  range, 
but  with  these  aural  courses  at  right  angles  to  the  visual.  This  facility,  therefore,  consists 
of  two  visual  and  two  aural  courses,  as  shown  in  Fig.  5.  Normally,  the  visual  course  is 


FIG.  5.     Basic  Patterns  and  Courses  for  Two-course  Visual  Range 

flown  by  observing  deviation  on  a  left-right  meter.    The  aural  signal  reverses  when  flying 
the  visual  course  across  the  station. 

The  electrical  system  of  the  visual  courses  is  identical  to  that  of  the  runway  localizer 
described  under  "Facilities  in  the  New  Federal  Airways  System"  in  article  4.  The  aural 
courses  are  laid  down  as  described  above  in  the  discussion  of  aural  radio  ranges.  Requiring 
less  course  sharpness  than  the  localizer,  its  antenna  array  consists  simply  of  three  horizontal 


RADIO  AIDS  IN  FEDERAL  AIRWAYS  SYSTEM  TODAY      22-09 


loops  for  the  visual  courses  and  two  additional  loops  for  the  aural.  The  central  loop  serves 
for  both  the  aural  and  visual  courses  (see  Fig.  6)  . 

A  simultaneous  voice  feature  is  incorporated  in  this  range  facility  by  modulating  the 
carrier  fed  to  the  center  loop  with  voice.  The  total  modulation  capacity  of  the  transmitter 
is  divided  approximately  as  follows:  90  cps  20  per  cent,  150  cps  20  per  cent,  1020  cps  10 
per  cent,  voice  40  to  50  per  cent. 

The  sharpness  of  the  visual  course  is  dependent  upon  the  transmitted  pattern  shape 
and  the  signal  voltages  delivered  to  the  receiver  indicating  circuit.  In  practice,  the 


Visual  loops 
North  South 


West 


FIG.  6.     Diagram  of  VHF  Two-course  Visual  Radio  Range 


receiver  gain  is  adjusted  to  give  full  scale  left  to  right  deflection  of  the  indicator  for  a  20° 
azimuthal  displacement  of  airplane  (plus  or  minus  10°  from  the  course).  The  aural  course 
sharpness  depends  upon  the  patterns  and  also  the  pilot  or  observer.  kOn  this  range  it  is 
about  2°. 

The  pilot  selects  the  voice  broadcasts  or  1020-cps  aural  signals  of  this  range  through 
the  standard  range-voice  filter  described  for  the  four-course  range. 

This  facility  is  subject  to  reflection  and  propagation  phenomena  characteristic  for  110 
Me  VHF.  Objects  near  the  station,  such  as  wires,  buildings,  and  trees,  reflect  the  signal 
and  give  multiple  path  transmission  to  the  receiver,  causing  scalloping  of  the  patterns 
and  sometimes  bends  or  multiple  courses.  In  a  moving  airplane  the  random  reflections 
of  trees  causes  wiggles  in  the  course  indicator.  Elevating  the  antenna  system  higher  to 
avoid  local  reflections  destroys  the  course  by  introducing  low-angle  nulls  in  the  vertical 
pattern  of  the  system.  The  conventional  counterpoise  (35-ft  diameter)  does  not  eliminate 
the  difficulty.  Horizontal  polarization,  originally  adopted  for  its  superior  performance 
in  the  instrument  landing  system  localizer,  aids  materially  in  reducing  the  reflection 
amplitude.  Probably  this  is  because  of  its  zero  radiation  at  the  horizontal  angle. 

Single  reflecting  sources  at  a  site  generally  give  smooth  (sinusoidal)  deviations  of  the 
course  indicator.  From  a  knowledge  of  the  wavelength  of  the  bends  and  the  position 
where  observed,  the  direction  of  the  reflection  source  can  be  determined  approximately 
by  calculation.  Distant  hills,  unless  extremely  elevated  above  the  station  and  observed 
by  a  low-flying  airplane,  do  not  cause  course  difiiculties.  Good  siting  is  important.  Station 
sites  must  be  fairly  flat  and  free  of  the  above  reflecting  sources  for  a  radius  of  about  500  ft. 
The  site  should  also  be  high  because  propagation  at  these  frequencies  is  line-of-sight 
(about  45  miles  is  realized  in  an  airplane  flying  1000  ft  above  the  station  elevation).  ^ 

RADIO  MARKERS  (75  Me).  An  essential  part  of  navigation  is  position  checking 
along  the  route.  The  radio  range  gives  only  direction,  or  lateral,  guidance.  The  null,  or 
"cone"  of  silence,  over  each  range  station  (Fig.  7)  has  been  used  for  years  as  a  means  of 


22-10 


AIDS  TO  NAVIGATION 


Direction  of  flight 


determining  "over  the  range"  position.  Intersecting  radio  range  legs  exist  at  some  places 
and  are  used  as  "fixes"  during  flight.  Special  non-directional,  low-power,  low-frequency 
markers  have  also  been  used.  The  difficulties  experienced  with  all  of  these  may  be  sum- 
marized as  follows:  (1)  unexpected  fade-out  of  signal  may  be  falsely  interpreted  as  the 
zone  over  the  station;  (2)  intersecting  range  legs  are  not  always  received  in  bad  weather; 

(3)  low-frequency  markers  have  been  heard 
many  miles  from  their  desired  area  at  night. 
The  development  of  the  positive  cone  or  Z 
marker  overcame  these  difficulties. 

The  Z  marker  consists  of  a  crystal-con- 
trolled, 3000-cps  tone-modulated,  75-Mc,  5- 
watt,  dual-transmitter  and  an  antenna  array. 

,-  _10i u  *    iji    x     Ll  'htoff_^j       The  transmitter  is  located  within  the  range 

te  fi    °n~7 j"| \J8|    °  station  and  is  a  dual  unit  with  automatic 

/  1  I  V— -"7  m.rb-,.  transfer  in  case  of  any  failure.  The  antenna 
is  located  between  the  radio  range  towers 
and  fed  from  the  main  station  building  by 


+20 


+10 


-20 


-30 


-40 
1.0 
Miles 

FIG.  7.     Range  Station  Cone  of  Silence  and  75-Mc 

Marker  Signal  Levels  Experienced  in  Flight  over 

Range  Station  at  1000-ft  Elevation 


transmission  line.  The  array  may  be  con- 
sidered as  two  pairs  of  horizontal,  collinear, 
half-wave  elements,  at  90°  to  each  other,  and 
excited  in  phase  quadrature.  The  array  is 
one-quarter  wavelength  above  a  mesh  coun- 
terpoise. The  resulting  service  from  this 
marker,  when  received  by  a  longitudinal 
doublet  under  the  airplane,  is  a  circular  pattern  (Fig.  4)  several  times  larger  than  the  cone 
of  silence  and  extending  upward  above  the  station  about  10,000  ft. 

The  receiver  generally  used  with  these  markers  is  a  crystal-controlled,  single-frequency, 
superheterodyne  unit  weighing  approximately  25  Ib  and  having  a  maximum  available 
sensitivity  of  about  150  microvolts.  Its  audio  output  is  filtered  and  connected  so  as  to 
cause  lighting  of  a  signal  lamp  on  the  airplane  instrument  panel  when  marker  signal  is 
received.  The  audio  output  is  connected  to  the  pilot's  headphone  circuit.  Three  different 
kinds  of  indicator  circuits  are  in  use  at  present  as  shown  in  Fig.  8.  The  simplest  is  that 
employing  a  rectifier  and  relay,  used  by  the  AAF.  One  white  light  is  connected  to  the 
relay  contacts.  The  CAA  and  airlines  use  a  receiver  with  no  relays  but  with  three  output 

Receiver  audio 
t 


Rectifier 


A.c.  from  Vibrator 
power  supply 

(a)  Army  (&)  Western  Electric  278 

FIG.  8.     Marker  Receiver  Signal  Circuits 


(c)  Bencflx 


audio  filters  and  three  different-colored  signal  lamps.  The  white  light,  operating  from 
3000-cycle  audio,  serves  for  the  airways  markers  (fan  and  #).  Amber  and  blue  lamps, 
operating  from  1300  and  400  cycles  respectively,  are  used  on  the  instrument  landing  system 
markers. 

Good  audio  selectivity  is  achieved  by  designing  the  indicator-reactor  circuit  to  operate 
at  about  0  db  and  by  limiting  the  receiver  audio  output  electronically  to  about  plus  6  db. 


RADIO  AIDS  IN  FEDERAL  AIRWAYS  SYSTEM  TODAY      22-11 


Any  undesired  signal,  being  able  to  produce  only  this  limited  output,  will  be  inadequate 
to  operate  the  lights  if  attenuated  merely  6  db.  Only  simple  filters  are  required. 

The  third  system  uses  the  audio  signal  directly  to  light  the  signal  light.  Good  filters 
are  employed.  When  any  desired  signal  appears  at  the  output,  some  of  it  is  rectified  and 
fed  back  to  increase  the  gain  of  a  controlled  amplifier  stage.  This  action  gives  regenerative 
response  resembling  relay  action. 

Horizontal  half-wave  doublet  antennas  are  generally  used  for  receiving  marker  signals. 
They  are  fastened  longitudinally  about  6  to  10  in.  under  the  belly  of  the  airplane.  Some- 
times the  same  antenna  is  also  coupled  to  the  low-frequency  range  receiver  so  as  to  serve  a 
dual  purpose.  Some  success  has  been  achieved  using  a  shortened  antenna  flush  with  the 
airplane  skin  and  having  a  reflecting  sheet  inside. 

Fan  marker  transmitting  stations  are  keyed,  tone-modulated,  100-watt,  75-Mc,  dual 
transmitters  connected  by  transmission  line  to  a  collinear  dipole  antenna  array  above  a 
mesh  counterpoise.  Three  thousand  cycle  tone  modulation  is  used,  and  it  is  keyed  in 
groups  of  dashes  to  identify  the  leg  of  the  range  on  which  it  is  located.  The  station  is 
located  on  or  near  the  center  of  the  range  course,  usually  about  20  miles  from  the  range 
station  (Fig.  4) .  Its  antenna  is  aligned  parallel  to  the  range  course  and  is  received  with 
greatest  efficiency  as  the  airplane  passes  over  or  to  either  side. 

The  name  "fan"  is  derived  from  the  shape  of  the  marker  field  pattern.  It  is  fan-shaped, 
as  shown  in  Fig.  9,  extending  perpendicularly  across  the  airway  so  as  to  be  received  even 


6000 


6 


6 


42024 

Distance  In  miles  from  station 
FIG.  9.     Typical  Dimensions  of  Airways  Fan-type  Marker 

by  aircraft  considerably  off-course.  Its  range  increases  with  altitude  but  is  generally 
receivable  6  miles. 'off  course  at  1000  ft  elevation. 

Most  of  the  257  fan  marker  stations  now  operating  have  an  antenna  system  using  four 
collinear  half-wave  elements  equally  spaced  180°  and  carrying  equal  currents  of  ^  the  same 
relative  phase.  The  array  is  one-quarter  wavelength  above  a  mesh  counterpoise.  Two 
minor  lobes  appear  in  the  radiation  pattern  in  the  area  directly  above  the  station.  A  new 
array  has  been  developed  and  is  being  installed  in  which  the  four  elements  carry  currents 
in  the  ratio  1-3-3-1  and  are  physically  spaced  220  electrical  degrees.  This  arrangement 
eliminates  the  minor  lobes  and  results  in  a  dumbbell-shaped  pattern  in  horizontal  cross- 
section. 

AUTOMATIC  DIRECTION  FINDER  (ADF)  RECEIVERS.  The  most  popular  and 
most  useful  radio  receiver  today  is  the  ADF  receiver  (Fig.  2).  It  is  an  all-purpose  receiver 
in  the  present  airways  system,  providing  for  the  reception  of  signals  from  radio  ranges, 
CAA  radio  communication  statiAis,  airport  control  towers,  and  airline  company  offices. 
This  receiver  operates  with  a  loop  and  a  short-wire  or  vertical  sense  antenna.  The  loop 
gives  accurate  directional  information  as  well  as  being  very  helpful  in  reducing  the  effects 
of  precipitation  and  thunderstorm  static. 

The  receiver  operates  as  an  ADF  by  virtue  of  the  introduction  of  an  audio  modulation 
into  the  loop  antenna  RF  circuit  as  shown  in  Fig.  10.  The  audio  frequency  is  non-critical 
and  is  usually  generated  in  the  receiver  at  about  48  cps.  The  loop  signal,  with  its  locally 
superimposed  modulation,  is  coupled  through  an  RF  transformer  to  the  non-directional 
antenna  circuit  and  to  the  IF  amplifier  circuit. 

When  the  loop  axis  coincides  with  the  direction  of  arrival  of  the  radio  waves,  the  loop 
contributes  no  RF  signal  to  the  receiver  system  and  consequently  none  of  the  48-cps  signal 
gets  through  to  the  receiver  IF  amplifier  or  audio  output.  When  the  loop  is  turned  so  as 
to  admit  signal,  the  polarity  (relative  phase)  of  the  48-cps  audio  output  with  respect  to 
the  original  48-cps  oscillator  output  is  dependent  upon  the  direction  in  which  the  loop  is 
turned  from  its  original  null.  A  pair  of  thyratron  tubes  in  a  balanced  modulator  circuit 
compares  these  phases  and  through  saturable  reactors  in  the  loop  motor  supply  circuit 
drives  the  loop  back  toward  the  null.  The  motor  and  control  circuit  causes  the  loop  to 
"seek"  and  hold  the  null  position  of  any  station  tuned  in  on  the  receiver.  A  selsyn  follow- 


22-12 


AIDS  TO  NAVIGATION 


up  system  is  used  to  indicate  remotely  to  the  pilot  the  position  of  the  loop.  The  loop 
position  indicator  is  calibrated  in  degrees  (0  to  360)  and  is  called  the  bearing  indicator. 
In  most  installations  there  is  some  distortion  of  the  received  waves  because  of  the  struc- 
ture of  the  airplane.  This  causes  an  error  in  indicated  bearing  called  "quadrantal  error." 
The  error  is  of  fixed  amount  in  given  directions,  and  the  amount  can  be  determined  by 
experiment.  Cam  systems  on  the  loop  shaft  are  usually  employed  to  compensate  for  the 
error  automatically. 

Pilots  peering 
Indicator  v 

X  i 

-  Indicator 


Loop 


Receiver 


Loop 

control 

equipment 


^Permanently  excited 
motor  winding 

FIG.  10.     Diagram  of  ADF  Receiver  Operation 

COMMUNICATIONS  (HP  AND  VHF).  All  communications  in  the  United  States 
airways  system  is  by  voice.  The  quality  of  voice  signals  received  from  aircraft,  and  some- 
times that  from  ground  stations,  is  far  below  any  standard  that  would  be  acceptable  in 
other  services.  As  air  traffic  increases,  there  will  be  increasing  need  for  quality,  or  for  a 
different  system,  to  preclude  misidentification  or  misunderstanding  of  received  calls.  This 
condition  is  not  easily  cured.  The  major  cause  of  poor  quality  is  the  enclosing  structure 
needed  around  the  microphone  to  exclude  the  100-db  noise  in  the  present-day  airplane 
cockpit.  The  ordinary  microphone  is  overmodulated  by  this  noise,  and  voice  cannot  be 
superimposed  without  severe  distortion.  The  aircraft  microphone  is  designed  to  be  rela- 
tively insensitive  by  enclosing  it  in  plastic.  High-level  voice,  obtained  by  close  talking, 
is  admitted  through  small  holes  in  the  plastic. 

Two  other  microphones  are  used.  One,  the  "throat  microphone,"  is  designed  to  pick 
up  vibrations  from  the  throat  by  being  worn  on  the  n^ck.  The  other  is  the  "lip  mic" 
operating  on  the  sound-velocity  principle.  Most  of  the  noise  pressures  strike  both  sides 
of  the  armature  of  this  mic  and  are  canceled.  By  attaching  it  to  the  pilot's  upper  lip, 
high-level  voice  is  applied  to  one  side  of  the  mic,  causing  modulation. 


Service 


Approximate 
Frequency  Band 


Long-range  navigation 70-200  kc 

Distress 500,  3105  kc,  121.5  Me 

Localizera 108-112  Me 

Radio  ranges 112-118  Me 

Air  traffic  control 118-122  Me 

Airline  communications 122—127  Me 

Air  to  ground 127-132  Mo 

Glide  path 329-335  Me 

Distance  measuring 960-1215  Me 

Radar  surveillance 3000  Me  band 

Precision  Radar 9090  Me 

FIG.  11.     Chart  of  Proposed  Frequencies  for  Air  Navigation  and  Communication 


FACILITIES  IN  THE   NEW  FEDERAL  AIKWAYS   SYSTEM      22-13 

Airborne  transmitters  generally  operate  in  the  3  to  6  Me  band  and  have  an  output  of  5 
to  100  watts.  The  lower-power  units  are  used  by  itinerant  fliers  in  contacting  traffic 
control  towers.  Plans  are  under  way  to  convert  all  communications  to  VHF.  Many 
ground  station  equipments  at  airport  towers  and  airway  stations  are  already  installed. 
The  VHF  bands  as  outlined  in  Fig.  11  will  permit  numerous  channels  of  static-free  serv- 
ice. The  aircraft  antenna  at  VHF  is  more  efficient  than  the  short  wires  currently  used, 
and  consequently  reliable  service  (not  beyond  line  of  sight)  can  be  expected  with  the 
same  or  less  transmitter  power.  All  airline  and  itinerant  transmitters  are  now  crystal 
controlled.  This  practice  will  continue  on  VHF  to  insure  reliable  service. 

AIR  TRAFFIC  CONTROL  AND  WEATHER  REPORTING.  The  air  traffic  control 
system  operated  by  CAA  on  the  federal  airways  depends  solely  on  voice  communication 
between  the  controller  and  the  pilot,  assisted  by  fan  markers  and  "holding"  markers 
about  which  planes  orbit  until  cleared  for  landing. 

Advanced  weather  information  is,  of  course,  essential.  Numerous  weather  stations 
now  release  radio-equipped  balloons  (radiosondes)  to  permit  study  of  the  upper  air  regard- 
less of  visibility  conditions.  These  radio  balloons  emit  coded  signals  revealing  altitude, 
humidity,  and  temperature  up  to  about  50,000  ft  altitude.  The  radio  equipment  is  ex- 
pendable but  is  protected  by  parachute  in  its  fall  to  earth;  some  are  picked  up  and  mailed 
back  for  reward.  Direction  finder  and  radar  tracking  of  radiosondes  has  permitted  deter- 
mination of  upper-air  velocity  and  direction. 

Two  new  electronic  devices  permit  automatic  measurement  and  recording  of  cloud- 
ceiling  height  and  horizontal  visibility,  the  "Ceilometer"  and  "Transmissometer"  re- 
spectively. 

The  Ceilometer  transmitter  uses  an  extremely  sharp  vertical  beam  of  high-intensity 
pulsing  light.  The  source  is  a  900-watt  mercury  arc,  striking  120  times  per  second.  At  a 
short  horizontal  distance  from  the  transmitter,  a  photoelectric  cell  scans  the  entire  beam 
from  its  base  to  top.  If  the  beam  is  striking  a  cloud  layer,  the  reflection  will  appear  in  the 
cell  output.  The  vertical  angle  of  cell  at  the  time  the  output  is  observed  is  indicative  of 
the  height  of  the  cloud  layer.  A  recording  device  makes  a  record  of  the  ceiling  altitudes. 
A  unique  feature  of  Ceilometer  is  the  use  of  pulsing  light  to  eliminate  the  otherwise  ob- 
literating effect  of  daylight.  Daylight,  being  steady,  is  filtered  out  of  the  system,  permit- 
ting equally  high  ceiling  measurement  performance  in  daytime  as  at  night. 

The  Transmissometer,  which  measures  the  light  transmissivity  of  the  air,  is  similar  to 
the  Ceilometer  in  utilizing  a  concentrated  beam  of  light  and  a  photoelectric  cell.  In  the 
Transmissometer,  however,  the  light  is  steady  and  directed  horizontally  to  the  cell  through 
a  kilometer  of  air  near  the  airport  approach  lane.  The  effect  of  daylight  is  removed  by 
shielding  and  baffling  and  by  proper  choice  of  beam  intensity.  The  output  of  the  cell  is 
converted  into  pulses,  the  lowest  pulses  corresponding  to  low  transmissivity.  In  the 
weather  bureau  these  pulses  are  converted  into  relative  values  of  direct  current  for  ob- 
servation or  recording  of  visibility. 

4.  FACILITIES  IN  THE  NEW  FEDERAL  AIRWAYS  SYSTEM 

The  new  federal  airways  system,  now  being  placed  in  operation,  will  utilize  VHF  radio 
for  ranges,  instrument  landing,  and  communications.  The  adoption  of  VHF  relieves  the 
troubles  of  static,  both  atmospheric  and  precipitation  types. 

The  new  radio  ranges  will  be  omnidirectional  to  satisfy  the  need  for  more  airways,  better 
traffic  control,  and  most  particularly  to  give  useful  navigation  information  regardless  of 
position  from  the  range  station. 

Distance-measuring  equipment  is  planned  as  an  ultimate  replacement  for  fan  and  Z- 
type  radio  markers.  Operating  with  the  omnirange  it  will  provide  the  basic  requirement 
for  safe  air  navigation — accurate  knowledge  of  position  at  all  tunes. 

Congestion  of  traffic  at  air  terminals  will  be  reduced  by  the  VHF  instrument  landing 
system  now  installed  and  by  closer  coordination  in  air  traffic  control. 

In  the  new  system  all  navigational  information  will  be  received  visually,  that  is,  by 
meter-type  presentation.  Some  use  of  cathode-ray  tubes  may  result  from  development 
work  now  under  way,  especially  for  anti-collision  and  air  traffic  control.  Whether  the 
presentation  is  by  meter  or  cathode-ray  tube,  some  effects  from  reflection  of  signal  by 
buildings,  trees,  or  mountains  near  the  ground  station  will  exist.  These  may  cause  bends, 
multiples,  or  wiggles  in  the  course  as  indicated  by  the  meter.  In  radar  systems,  wherein 
all  the  intelligence  is  obtained  by  visual  study  of  the  cathode-ray  tube,  the  operator  can 
generally  separate  the  main  from  the  reflected  signal  visually.  Pulse  technique  does  not 
in  itself  eliminate  the  effects  of  reflections,  but  when  displayed  against  a  time  base^  on  a 
cathode-ray  tube  it  permits  a  study  of  all  signals  received.  The  reflected  signals  are  visibly 
displaced  and  diminished  by  the  extra  tune  required  to  travel  their  longer  paths* 


22-14 


AIDS  TO  NAVIGATION 


INSTRUMENT  LANDING  SYSTEM  (CAA).  Instrument  approach,  or  landing  by 
instrument  guidance,  is  just  now  being  put  into  practice  in  the  United  States.  It  comprises 
a  runway  localizer,  a  glide  path,  and  three  marker  beacons. 


FIG.  12. 


|90<M| 
Alt.  Motor  Alt. 

Diagram  of  CAA  Electronically  Modulated  Localizer 


Localizer.  The  localizer  creates  a  course,  or  track,  along  the  center  line  of  the  runway, 
by  overlapping  two  bean-shaped  radio  patterns  having  different  modulation  frequencies 
(90  and  150  cps).  The  signal  service  area  extends  slightly  beyond  the  line  of  sight,  but  is 

conservatively  given   as  25  miles  at 

|  1000  ft  (airplane  elevation).    The  fre- 

J  quency  is  110  Me. 

The  localizer  transmitting  antenna 
array  consists  of  eight  horizontally 
polarized  loops,  spaced  symmetrically 
across  the  center-line  extension  of  the 
runway.  It  is  generally  1500  ft  from 
the  end  of  the  runway  and  elevated 
only  about  12  ft.  Each  of  the  three 
outside  loops  on  one  side  is  paired 
electrically  with  a  loop  in  similar  po- 
sition on  the  other  side.  Each  pair  is 
connected  through  a  control  to  a  com- 
mon sideband  generator,  as  shown  in 
Fig.  12.  There  is  a  180°  phase  re- 
versal in  the  tie  line  between  the  pairs. 
The  sideband  energy  is  radiated  with 
a  sharp  null  down  along  the  runway 
center  line,  as  in  Fig.  13a.  The  center 
two  loops  are  connected  in  phase 
(Fig.  12)  to  radiate  carrier,  modulated 
with  equal  amounts  of  90-  and  150-cps 
voltage.  The  radiation  pattern  is 
shown  in  Fig.  136.  When  the  side- 
bands in  patterns  a  and  b  of  Fig.  13 
combine  in  the  receiver  and  appear 
FIG.  13(o).  Antenna  Pattern  of  the  CAA  8-loop  Local-  at  the  filtered  output,  the_  overlapping 


izer.     Side-band  loops. 


patterns  of  Fig.  13c  are  obtained. 


FACILITIES  IN  THE  NEW  FEDERAL  AIRWAYS  SYSTEM      22-15 


The  airplane  receiver,  Fig.  14,  is  adjusted  initially  so  that  its  filtered  output  of  separate 
90-  and  150-cps  voltage  is  well  balanced.  The  output  is  connected  to  a  balanced  rectifier, 
and  the  rectifier  is  connected  to  a  zero-center  microammeter.  The  meter  remains  centered 
as  long  as  equal  amounts  of  90-  and 

150-cps  modulation  are  received,  as  ^ 

when  flying  along  the  runway  center  | 

line  in  the  sideband  antenna  null. 
Deviation  of  the  airplane  right  or 
left  brings  it  out  of  the  null.  The 
sideband  signal  then  received  adds 
to  that  obtained  from,  the  carrier. 
On  one  side  of  the  null  (or  course) 
the  90-cps  signal  adds  to  that  of  the 
carrier  while  the  150-cps  signal  sub- 
tracts. This  causes  deflection  of  the 
zero-center  meter.  The  reverse  is 
true  on  the  opposite  side  of  the 
course.  The  zero-center  meter  (Fig. 
15)  is  used  by  the  pilot  to  determine 
deviation  from  and  ^direction  to  the 
course.  The  same  instrument  usu- 
ally contains  a  second  pointer,  cen- 
tered horizontally  to  indicate  devia- 
tion from  the  glide  path.  The  "U" 
receiving  antenna  is  shown  in 
Fig.  21a. 

Although  it  is  possible  to  obtain 
almost  unlimited  course  sharpness 
in  the  localizer  course  by  expanding 
the  transmitting  array  and  increas- 
ing the  receiver  output  level,  there 
is  a  maximum  that  can  be  used 
safely  and  conveniently  by  average 

pilots.     The  sharpness  will  probably  be  standardized  at  5 
right,  for  2.5°  deviation  from  course). 

Only  part  of  the  200-watt  carrier  (40  per  cent)  radiated  by  the  center  pair  of  loops  is 

modulated  by  the  90/150  cps  signals. 

S*  The  balance  is  used  to  handle  tone 

identification  keying  and  control  tower 
voice  modulation.  The  latter  is  ex- 
pected to  be  an  aid  in  approach  control 
of  airport  traffic. 

Glide  Path.  The  glide  path  is 
established  by  a  crystal-controlled 
transmitter  capable  of  delivering 
about  25  watts  at  330  Me  to  an  an- 
tenna system  producing  two  types  of 
lobes.  One  antenna  produces  a  beam 
quite  broad  in  the  vertical  plane  (see 
Fig.  16a)  which  is  modulated  at  90 
cycles.  This  will  be  called  the  upper 
beam.  The  other  antenna  is  modu- 
lated at  150  cycles  and  is  raised  sev- 
eral wavelengths  off  the  ground  to 
produce  a  multilobed  pattern  with 
each  lobe  quite  narrow  in  the  vertical 
plane.  Comparison  of  the  broad  up- 
per lobe  with  the  lowest  of  the  narrow 
lobes  establishes  the  glide  path.  This 
equisignal  path  is  generally  about  2.5°, 
although  adjustment  of  the  heights  of 
the  antennas  can  bring  the  crossover 
at  any  angle  between  2°  and  4°.  An 
accurate  plot  of  the  lobes  involved 
FIG.  13  (c).  Combined  Side-band  Pattern  (a)  and  (6)  for  the  2.5°  path  is  shown  in  Fig.  17. 


FIG.  13(6).    Antenna  Pattern  of  the  CAA  8-loop  Local- 
izer.    Carrier  loops. 


(deflection  to  last  dot  left,  or 


22-16 


AIDS  TO  NAVIGATION 


The  separate  modulation  frequencies  are  generated  by  a  mechanical  modulator  consist- 
ing of  a  synchronous  (1800-rpm)  motor  and  two  metal  paddle  wheels  having  three  and 
five  paddles,  respectively.  The  paddles  detune  associated  resonant  sections  of  trans- 
mission line  coupled  to  the  respective  antennas  and  create  100  per  cent  modulation  at  90 
and  150  cps.  The  modulator  is  shown  in  the  diagram,  Fig.  166. 

±2 


Fll'ters 


Horrzontal  pofnteY 
used  only  with  ils 
glide  path  receiver 


Course 

sharpness 

control 


§< h^/WW  )  < 


FIG.  14.     Localizer  Receiver  Output  Circuit 


.Pilot's  cross 
pointer  Instrument 


The  equisignal  surface  represented  by  the  two  radiations  of  the  station  is  obviously 
conical;  paths  from  all  directions  terminate  at  the  station.  The  station  had  to  be  placed 
off  the  side  of  the  runway  for  safety  reasons.  This  means  that  the  approaching  airplane, 
if  in  line  with  the  runway,  must  follow  a  hyperbolic  path  whose  minimum  altitude  exists 
directly  opposite  the  station.  To  straighten  out  the  bottom  portion  a  special  relation 
between  the  horizontal  patterns  of  the  upper  and  lower  antennas  had  to  be  applied. 

The  receiver  for  the  glide  path  uses  a  crystal-controlled  superheterodyne  circuit  and  a 
28-volt  d-c  supply.  It  requires  no  high-voltage  dynamotor  or  vibrator  power  supply. 


cr 


o 


(6) 

FIG.  15.     Cross-pointer  Instrument  with.  Flag-alarms  to  Indicate  Signal  Failure,     (a)   Dead  needle 
position  showing  alarm  flags  in  position.     (6)  Operative  position  with  alarm  flags  behind  mask.     (Cour- 
^  tesy  AAF.) 

The  output  audio  (90  and  150  cps)  is  filtered  and  separately  rectified  as  in  the  localizer 
receiver,  Fig.  14.  The  resultant  direct  current  is  connected  to  a  zero-center  instrument 
(horizontal  pointer  of  cross  pointer  instrument,  Fig.  15).  The  pilot  flies  on  the  path, 
keeping  the  pointer  centered,  or  horizontal.  The  dipole  antenna,  shown  in  Fig.  21a,  is 
used  hi  receiving  the  glide-path  signals. 


FACILITIES  IN  THE  NEW  FEDERAL  AIRWAYS  SYSTEM      22-17 


It  is  evident  in  that  the  glide-path  equisignal  lines  converge  and  become  very  sensitive 
to  vertical  displacement  of  the  airplane  near  the  station  and  runway.  This  is  offset  by 
designing  into  the  receiver  about  9  db  of  negative  AVC.  This  has  the  effect  of  reducing 
the  receiver  audio  output  voltage  as  the  airplane  approaches  the  station.  An  additional 


Upper  ant. 
150  ™ 


Lower  ant. 
90f\j 


Coupled  section 


Mechanical 
modulator 


•Transposition 

^ntl-cross  modulation 
bridge 

M  =  motor 

L  =  load  balance 


90  ru  pattern 
"(lower  ant.) 


Ground 


(W 


V>*/  ^uj 

FIG.  16.     333-Mc  Straight-line  Equisignal  Glide  Path,     (a)  The  lobes  of  elevated  antennas  produce 
an  equisignal  glide  path.     (6)  Modulation  is  accomplished  mechanically  with  coupled  sections. 

4-db  reduction  or  "softening"  of  sharpness  is  derived  by  beaming  unmodulated  carrier 
across  the  path  near  its  bottom  end. 

Markers.  The  markers  used  in  instrument  landing  are,  fundamentally,  position- 
indicating  devices  similar  to  the  Z  markers  discussed  on  p.  22-10.  They  will  be  replaced 
eventually  with  accurate  distance-measuring  devices  which  continuously  indicate  mileage 
to  the  runway.  The  present  two  markers  are  identified  both  aurally  and  visually  by 
tone  and  keying.  The  outer  marker  (distance  4.5  miles)  is  modulated  at  400  cps  and 
keyed  in  long  (2  per  second)  dashes.  The  middle  marker  (distance  3500  ft)  is  modulated 


1 

J 

& 

c 

, 

x^ 

^^ 

X 

/ 

' 

\ 

/ 

f 

ver  radiator 
(565.3°  sin 

a) 

\ 

sin 

2.5° 
path 

/ 

\ 

16.5° 
False  path  with 
reverse  sensing 

/ 

. 

/  Up 
/0.4 

62  sir 

iator 
(268 

5°  sin 

a) 

s 

^ 

1 

I 

r 

/ 

r 

^ 

^ 

, 

s^ 

\ 

I 

^ 

1 

\ 

/ 

\ 

/ 

\ 

/ 

\ 

Y 

/ 

/ 

\ 

/ 

\ 

/ 

\ 

/ 

\ 

n 

\ 

// 

\ 

/ 

\ 

/ 

\ 

\/ 

'  i 

i 

\ 

L 

\ 

w 

V 

V 

i 

D         1        2        3        4        5        6        7        8        9        10      11       12       13      14      15      16      17      1« 
Vertical  angle  a 
FIG.  17.    Accurate  Plot  of  Radiations  Showing  Formation  of  Glide  Path 

at  1300  cps  and  is  keyed  in  a  series  of  dot-dash  characters.  The  boundary  marker,  re- 
cently discontinued,  was  modulated  at  3000  cps  and  was  keyed  in  fast  dots  (6  per  second). 
The  outer  and  middle  markers  are  further  identified  by  signal  lamp  color,  purple  and  am- 
ber respectively. 


22-18 


AIDS  TO  NAVIGATION 


THE  OMNIDIRECTIONAL  RANGE  (VHF).  This  facility  is  produced  by  radiating 
simultaneously,  from  the  same  antenna  array,  two  signals  having  the  same  audio-frequency 
modulation  but  having  different  relative  phase  for  different  azimuthal  positions  around 
the  array.  One  signal  is  non-directional,  and  since  its  phase  is  everywhere  (in  azimuth) 
the  same  it  is  called  the  "reference  phase"  or  voltage.  The  second  is  produced  by  rotating 
a  "figure-of-eight"  pattern.  This  pattern  produces  a  modulation  in  the  radio  receiver, 
the  frequency  being  dependent  on  the  speed  of  rotation.  Since  its  relative  phase  varies 
with  azimuth  it  is  called  the  "variable  phase"  or  voltage.  Since  the  two  frequencies  gen- 
erated are  equal,  and  a  standard  relation  is  set  for  north,  comparison  of  the  relative  phase 
anywhere  will  permit  determination  of  azimuth  bearing  from  the  station. 

A  five-loop  transmitting  array  is  used,  and  it  is  mounted  on  a  tower  15  ft  high  with  a 
circular  counterpoise.  Four  loops  are  installed  as  diagonally  opposite  pairs.  These  are 
connected  (Fig.  18)  to  the  transmitter  through  a  capacitance  goniometer  whose  rotation 


I  30<\>  reference 


FIG.  18.     The  CAA  VHF  Omnidirectional  Range  System  Diagram 

at  1800  rpm  causes  the  figure-of-eight  space  pattern  of  the  loops  to  rotate  in  synchronism. 
It  produces  a  30-cps  modulation  of  the  carrier  in  the  receiver. 

The  center  antenna  is  connected  directly  to  the  transmitter.  The  transmitter  is  modu- 
lated by  a  30-cps  voltage  derived  from  a  tone-wheel  generator  locked  on  the  goniometer 
shaft.  If  the  two  30-cps  signals  in  the  receiver  output  are  in  phase  when  the  plane  is  north 
of  the  station,  the  phase  difference  in  degrees  elsewhere  agrees  directly  with  the  degrees  in 
azimuth  of  the  receiver. 

Two  30-cps  signals  cannot  be  directly  isolated  in  the  transmitter  or  receiver,  so  a  sub- 
carrier  of  10  kc  must  be  used  for  the  reference  signal.  Further  isolation  is  provided  by 
frequency-modulating  the  subcarrier  with  the  30-cps  voltage.  Both  the  subcarrier  and 
its  required  FM  are  derived  from  the  tone-wheel  pickup  by  using  non-uniform  spacing  of 
teeth  on  the  wheel.  A  "clipper"  (see  Fig.  18)  is  used  to  remove  modulation  on  the  RF 
to  the  goniometer. 

The  rotating  pattern,  which  is  sideband  energy,  effectively  modulates  the  carrier  about 
30  per  cent.  The  subcarrier  modulation  is  equivalent  to  10  per  cent.  The  remainder  of 
carrier  is  utilized  for  tone-keyed  identification  and  voice  communications. 

The  receiver  used  is  a  conventional  superheterodyne  up  to  the  second  detector.  Here  a 
filter  is  imposed  to  separate  the  10-kc  subcarrier  and  the  variable  30-cps  modulation.  The 
10-kc  subcarrier  is  fed  to  a  discriminator  from  which  the  reference  30-eps  signal  is  obtained 
as  illustrated  in  Fig.  18. 

The  two  30-cps  receiver  output  signals  are  connected  to  a  wattmeter  circuit.  The  indi- 
cating element  of  the  wattmeter  is  a  zero-center  instrument  which  is  used  as  a  course 
indicator.  When  the  phase  difference  between  the  reference  and  variable  signals  delivered 
to  the  circuit  is  90°,  the  indicating  instrument  will  be  centered.  This  may  be  standardized 
as  the  north,  or  east,  bearings  of  the  station.  Varying  of  the  receiving  point  from  this 


FACILITIES  IN  THE  NEW  FEDERAL  AIRWAYS  SYSTEM      22-19 


bearing  will  cause  movement  of  the  zero-center  meter.  Full-scale  reading  is  obtained  in 
the  system  when  the  receiving  point  is  displaced  10°  from  north.  A  variable  phaser, 
Fig.  18,  is  provided  to  delay,  or  advance,  the  one  signal  before  delivery  to  the  wattmeter 


Knob 


Azimuth 
selector 
(phaser) 


Flag  (Signal  failure)- 
alarm 


(a)  Circuit  for  Manual  Selection  of  Omnirange  Track 
Filter  Ampl'.  Discriminator        I        ^ff       |  Amp!'. 


10  kc 


Magnetic  vane 
Automatic  &  pointer 

bearing 
Indicator 


Audio  output 


+  B 


(6)  Circuit  for  Omnirange  Automatic  Bearing  Indicator 
FIG.  19.     Aircraft  Receiver  Output  Indicating  Circuits 

circuit.  If  sufficient  phase  change  is  inserted  to  center  the  pointer  of  the  indicating  instru- 
ment, the  amount  of  phase  change  inserted  is  proportional  to  the  azimuth  bearing.  The 
phaser  is  calibrated  0°  to  360°.  This  phaser  setting  is  taken  as  the  airplane-to-station 
bearing  when  the 'zero-center  meter  pointer  is  centered. 

Obviously,  there-  are  two  positions  of  the  360°  phaser  where  the  indicating  instrument 
will  be  centered.    This  ambiguity  is  indicated  either  by  a  combination  of  red  and  green 


22-20  AIDS  TO  NAVIGATION 

lights  or  by  a  smaller  zero-center  ambiguity  meter.  When  signal  lights  are  used,  they  are 
operated  from  a  relay  which  is  normally  closed  for  correct  signal  and  open  for  incorrect, 
or  insufficient,  signal.  The  relay  closing  is  achieved  by  combining  the  two  voltages  in 
phase  through  an  amplifier.  Since  the  voltages  are  normally  equal,  reversal  of  one  causes 
complete  cancellation,  the  relay  releases,  and  the  red  light  comes  on. 

The  alternative  method  uses  a  wattmeter  circuit  similar  to  that  in  the  original  course 
indication;  however,  its  variable  signal  is  shifted  90°.  It,  therefore,  indicates  a  course  90° 
displaced  to  the  true  course.  The  ambiguity  meter  indicates  full  scale  right  until  the 
airplane  passes  over  the  station.  At  this  instant  it  swings  to  the  opposite  side,  indicating 
that  the  station  has  been  passed  and  that  the  bearing  is  reciprocal.  When  the  ambiguity 
meter  centers,  it  indicates  failure  of  one  or  both  signals  or  an  inoperative  receiver.  One 
such  meter  is  diagrammed  in  Fig.  19a. 

Figure  196  illustrates  an  automatic  "bearing  indicator"  for  the  omnirange.  Its  pointer 
follows  up  the  bearing  changes  made  by  the  airplane,  always  displaying  correct  bearing. 
It  is  a  magnetic  device  with  crossed  coils  connected  to  the  respective  rectified  variable 
and  reference  phases  from  the  receiver.  Cathode-ray  tube  and  other  types  of  indicators 
have  been  developed.  One  system  combines  the  omni-  and  the  gyrosyn-magnetic  com- 
pass to  give  the  equivalent  of  VHF  ADF. 

DISTANCE-MEASURING  EQUIPMENT.  The  distance-measuring  equipment  with 
meter  indication  depends  for  its  operation  on  the  challenging  of  a  ground  "radar  trans- 
pondor  beacon"  by  an  airborne  challenger.  The  challenger  is  a  radio  transmitter-receiver 
which  transmits  a  pulse-type  challenge  to  and  receives  a  pulse-type  reply  from  an  auto- 
matic radio  receiver-transmitter  known  as  a  transponder  beacon.  The  time  elapsed 
between  transmission  of  the  challenge  and  receipt  of  the  reply  is  a  measure  of  the  distance 
between  the  challenger  and  the  beacon.  This  time  difference  may  be  converted  to  a  d-c 
voltage  whose  magnitude  is  proportional  to  distance  and  can  be  read  on  a  meter. 

The  challenge  signal  consists  of  a  "pair"  of  RF  pulses  in  the  960-990  Me  band  so  that  a 
beacon  setting  must  agree  with  two  pulse  characteristics  before  the  beacon  replies.  These 
are  the  challenge  frequency  and  the  time  separation  (called  mode)  between  the  two  pulses 
of  a  pair.  This  "frequency-mode"  combination  is  called  a  challenge  channel.  The  reply 
channel  consists  of  a  "pair"  of  RF  pulses  in  the  1185-1215  Me  band.  Challenge  and  reply 
channels  in  combination  form  an  "operating  channel."  The  beacon  and  the  challenger 
receiver  distinguish  between  pulses  intended  for  them  and  pulses  at  the  same  frequency 
but  different  modes  by  use  of  a  "decoder."  One  form  of  decoding  is  to  delay  the  first 
pulse  by  an  amount  equal  to  the  mode  and  operate  only  if  the  delayed  pulse  is  then  co- 
incident with  the  second  pulse. 

An  individual  plane  identifies  the  replies  to  its  own  challenges,  as  distinguished  from 
replies  of  the  same  beacon  to  the  challenges  of  other  planes,  by  the  fact  that  its  own  replies 
occur  after  a  fixed  (or  slowly  changing)  time  (i.e.,  distance)  after  its  challenge,  while  the 
replies  to  other  planes  occur  at  a  random  time.  The  airborne  receiver  is  "gated  on"  for  a 
very  short  time  interval  at  a  fixed  time  (i.e.,  distance)  after  each  challenge.  It  is  as  if  the 
receiver  asked  the  question,  "Is  there  a  reply  from  a  beacon  at  a  distance  between  20  and 
22  miles?"  The  gate  duration  is  defined  as  22  minus  20  or  2  miles.  In  one  design  the  gate 
is  divided  into  two  subgates  called  (1)  the  "early  gate"  and  (2)  the  "late  gate."  If  the 
signal  occurs  in  the  early  gate,  the  time  delay  of  the  gate  is  decreased  for  later  challenges 
(it  is  gated  on  at  19  instead  of  20  miles).  Likewise,  if  the  signal  occurs  in  the  late  gate, 
the  time  delay  is  increased  for  later  challenges  (it  is  gated  on  at  21  instead  of  20  miles). 
This  system  of  subgates  allows  automatic  following  of  the  signal.  The  time  between  the 
challenge  and  the  "gate"  is  translated  into  volts  by  a  measuring  sawtooth  voltage  which 
defines  the  volt-time  (i.e.,  volt-distance)  relationship. 

While  searching  for  a  beacon,  the  gate  is  made  to  travel  slowly  over  the  total  distance, 
advancing  a  fraction  of  its  duration  after  each  challenge.  It  is  as  if  the  gate  asked  the 
question,  "Is  there  a  reply  between  20  and  22  miles?"  then  after  the  next  challenge  it 
asked,  "Is  there  a  reply  between  20.5  and  22.5  miles?"  etc.  If  the  gate  finds  a  high  enough 
percentage  of  replies  to  challenges,  the  search  is  terminated  and  tracking  begins;  that  is, 
the  receiver  thereafter  holds  the  signal  which  has  been  found. 

Each  beacon  is  identified  by  "gap  coding";  i.e.,  its  transmission  is  interrupted  by  a 
keyer  in  such  a  manner  that  the  gaps  form  a  Morse  character.  Failure  to  receive  a  reply 
results  in  lighting  a  lamp  in  the  cockpit  which  therefore  flashes  the  beacon's  code. 

A  beacon  is  disabled  for  a  short  time  after  replying.  This  time  is  called  '  'beacon  recovery 
time"  or  "beacon  dead  time."  While  this  "recovery  time"  may  serve  many  useful  purposes 
such  as  preventing  over-interrogation,  it  also  results  in  failure  to  reply  to  those  challenges 
which  occur  while  the  beacon  is  recovering  from  a  previous  interrogation.  This  failure 
is  called  "count-down"  and  is  expressed  as  the  ratio  of  missed  replies  to  the  total  number 
of  challenges. 


FACILITIES  IN   THE  NEW  FEDERAL  AIRWAYS  SYSTEM      22-21 


The  number  of  aircraft  signals  that  can  be  handled  simultaneously  by  a  given  beacon, 
is  very  important.  The  system  described  above,  through  choice  of  pulse  dimensions  and 
random  rate,  can  handle  approximately  fifty  airplanes  per  channel  simultaneously  (sixty 
channels  are  repeated  in  every  500  mile  square) .  A  block  diagram  of  the  airborne  unit  is 
shown  in  Fig.  20. 

Distance  can  also  be  measured  by  the  same  principle  as  that  employed  in  the  radio 
altimeter,  but  with  the  aid  of  ground  respondor  stations.  Continuous-wave  frequency- 
modulated  transmission  is  employed.  The  modulation  is  varied  over  a  cycle  which  is, 
in  duration,  equal  to  at  least  the  time  required  for  travel  of  the  signal  to  and  from  the 
most  distant  beacon  to  be  measured.  The  frequency  of  the  received  signal  at  any  instant 
is  different  from  that  then  being  emitted.  The  difference  is  a  function  of  the  time  and 
therefore  the  distance.  The  meter-style  indicator  can  be  calibrated  in  miles.  The  disad- 
vantage of  this  CW  system  is  that  it  can  handle  only  one  aircraft  at  a  time. 


Trans- 

Mod. 

Multivlb. 

Linear 
sweep 

mitter 

Mileage 
Indicator 
(voltmeter^ 


FIG.  20.     Functional  Diagram  of  Distance-measuring  Equipment  (Airborne  Unit) 

Audio  phase  comparison  methods  have  been  proposed  and  tried  experimentally.  In 
this  method  an  audio  modulation  frequency  is  chosen  whose  wave  period  (360°)  is  the 
same  as  the  time  required  for  transmission  to  and  from  the  maximum  distance  to  be 
measured.  The  radio  signal  travels  186  miles  in  1  ms.  Therefore,  a  1000-cps  frequency 
will  shift  360°  in  186  miles.  If  the  airplane  and  ground  station  are  93  miles  apart,  the 
round  trip  between  them  is  186  miles  and  the  360°  shift  will  prevail  as  the  maximum  range 
for  the  1000-cps  wave. 

By  sharing  time  through  random  keying  of  the  airborne  transmitters,  several  aircraft 
may  obtain  separate  distance  information  simultaneously  in-  the  audio  phase  comparison 
system  from  the  same  ground  station. 

AIRCRAFT  EQUIPMENT  (RECEIVING).  Almost  universally,  aircraft  receivers  have 
been  of  the  superheterodyne  type.  Tunable  receivers  have  been  used  generally,  with  spot 
frequency  for  control  towers  (278  kc)  crystal-controlled  or  fix-tuned.  In  the  VHP  system 
the  superheterodyne  principle  is  again  being  universally  used.  Crystal  control  is  benig 
used  for  most  commercial  .applications  and  manual  tuning  for  inexpensive  units  for  itin- 
erant fliers.  In  the  crystal-controlled  receivers  each  of  the  several  crystals  employed  will 
produce  a  multiplicity  of  receiving  channels.  Over  two  hundred  channels  are  needed  by 
aircraft  on  routine  instrument  nights,  in  the  bands  illustrated  in  the  national  plan,  Fig.  11. 

Many  companies  have,  or  are  installing,  dual  ADF  receiving  equipments  in  each  air- 
plane so  that  automatic  indication,  or  plotting,  of  aircraft  position  can  be  used.  Single 
instruments  with  dual  pointers,  one  for  each  ADF  receiver,  and  each  receiver  tuned  to  a 
separate  station,  are  very  helpful  in  navigation.  ^ 

VHF  receivers  for  navigation  are  equipped  to  serve  all  VHF  functions  required  in  flagkt. 
Two  such  receivers  in  an  airplane  give  adequate  stand-by  protection.  In  normal  use  one 
is  available  for  communication  while  the  other  is  being  used  for  navigation  or  landing. 

Two  types  of  output  indicating  circuits  are  now  used  in  each  of  these  navigation  re- 
ceivers. The  amplitude  comparison  type  (90-150  cps)  will  be  retained  only  until  existing 
localizers  of  the  instrument  landing  systems  are  converted  from  amplitude  to  piiase 
comparison  principle.  Then  the  phase  comparison  type  indicating  circuit  will  be  used 
for  both  omni  navigation  and  landing.  m  . 

An  outstanding  advancement  in  receivers  was  made  during  the  war  in  the  elimination 
of  aU  receiver  high-voltage  power  supply.  Through  the  development  and  application  of 
the  28-volt-type  tube  (28D7)  a  complete  330-Mc,  superheterodyne,  glide  path  receiver, 
using  no  vibrator  or  dynamotor-type  supply,  was  produced  in  quantity.  - 


22-22  AIDS  TO  NAVIGATION 

volts  direct  current,  available  in  the  airplane,  is  used  as  plate  supply.  An  ADF  receiver 
was  developed  later,  also  avoiding  the  HV  supply.  The  reduction  in  noise  and  main- 
tenance and  the  saving  in  weight  by  this  development  are  extremely  important  factors. 
Receiving  antennas  have  been  an  important  part  of  the  VHF  program.  Whip  or  mast- 
type  vertical  antennas  are  the  best  for  ease  of  installation.  For  lateral  guidance  functions, 
however,  horizontal  polarization  has  been  declared  superior.  Suitable  antennas  are  illus- 
trated in  Fig.  21.  The  patterns  of  the  U  and  V  antennas  are  essentially  circular,  as  required 
for  their  navigational  function  in  range  and  localizer  receptions.  The  330-Mc  dipole  on 
the  U  antenna,  in  Fig.  21,  is  used  for  glide  path  reception.  It  is  required  to  have  only 
forward  glide  path  reception  during  the  runway  approach  procedure. 


330  me  glide  path 

110  me  localizer 


(a) 


FIG.  21.     Receiving  Antenna  for  the  Instrument  Landing  System,     (o)  The  U  is  for  110-Mc  band 
localizers,  and  the  dipole  is  for  the  330-Mc  glide  path.     (6)  An  experimental  V  antenna  for  non- 
directional  reception  of  localizer  or  range  signals. 

For  navigational  service,  pure  polarization  is  essential  in  the  radiated  wave.  If  polariza- 
tion is  impure,  that  is,  unintentionally  mixed,  serious  attitude  effects  are  evident  in  the 
indicator  when  the  airplane  is  banked  or  turned. 

Balanced  transmitting  and  receiving  antennas  are  now  used. 

AUTOMATIC  FLIGHT  AND  LANDING  EQUIPMENT.  For  automatic  navigation, 
the  new  electric  auto-pilot  has  become  the  greatest  asset.  With  it,  straight  and  level 
flight  is  made  by  gyroelectric  control,  or  completely  coordinated  (aerodynamically)  turns 
of  the  airplane  may  be  made  by  turning  a  knob  which  simply  unbalances  electrical  bridges 
in  the  auto-pilot  system.  If  the  bridge  circuit  is  electrically  connected  to  the  output  of 
the  new  navigation  receiver  (localizer  or  radio  range)  through  an  appropriate  amplifier 
or  coupling  system,  the  auto-pilot  can  be  made  to  fly  the  airplane  accurately  along  the 
localizer  or  radio  range  course.  Or  the  output  of  the  receiver  (direct  current  which  is 
proportional  to  displacement  from  the  course  and  whose  polarity  reverses  when  crossing 
the  course)  may  be  made  to  operate  a  steering  motor  right  or  left,  to  follow  the  course. 
The  steering  motor  turns  the  auto-pilot  steering  potentiometer. 

The  operation  of  one  type  of  electrical  auto-pilot,  and  one  way  in  which  the  radio  signal 
may  be  coupled  to  the  auto-pilot,  are  shown  diagrammatically  in  Fig.  22. 

Coupling  the  radio  guiding  signal  to  the  auto-pilot  and  obtaining  satisfactory  perform- 
ance involves  consideration  of  the  mass  and  speed  of  the  airplane,  sharpness  of  the  radio 
course,  and  characteristics  of  the  auto-pilot  itself.  The  desired 'performance  is  that  giving 
asymptotic  approach  to  the  course.  In  the  off-course  position,  a  "displacement"  signal 
must  be  applied  to  turn  the  airplane  right  or  left  as  required.  The  displacement  signal  is 
obtained  directly  from  the  radio  receiver  and  is  proportional  to  the  angular  distance  from 
the  course.  Acting  alone,  this  displacement  signal  would  reduce  the  turn  of  the  airplane 
to  zero  as  it  crosses  the  course.  But  the  airplane  heading  at  the  time  of  crossing  may  be 
at  any  angle  to  the  desired  course.  The  airplane  travels  to  the  opposite  side  of  the  course 
before  a  reverse  signal  is  applied.  The  result  of  displacement  signal  alone  would  be  a 
continuous  oscillation  of  the  airplane  across  the  course  as  it  flies  toward  the  guiding  radio 
station. 

In  the  coupling  device,  a  "rate"  signal  must  be  applied — a  signal  whose  amplitude  is 
proportional  to  the  rate  of  change  of  displacement,  and  whose  control  on  the  airplane 
through  the  auto-pilot  is  reverse  from  that  of  the  displacement  signal.  With  proper  design 
for  any  particular  airplane,  the  rate  circuit  reverses  the  turn  of  the  airplane  as  it  nears 
the  course  and  causes  it  to  follow  the  desired  asymptotic  curve.  Because  of  the  converging 
sharpness  feature  of  radio  courses,  a  compromise  must  be  made  in  the  coupling  system 
characteristics.  In  general,  however,  good  performance  has  been  achieved.  On  the 
final  approach  to  the  airport  on  the  localizer,  the  vertical  pointer  of  the  course  indicator 
seldom  deviates  from  center  by  more  than  its  own  width,  regardless  of  any  cross-wind 
velocity  or  direction. 


FACILITIES  IN  THE  NEW  FEDERAL  AIRWAYS  SYSTEM      22-23 


Flight  toward  or  away  from  the  station  is  obtained  by  reversal  of  connections  (polarity) 
between  the  radio  receiver  and  coupling  unit.  The  switch  positions  are  marked  "inbound" 
and  "outbound"  respectively. 

Very  flexible  radio  range  navigation  is  now  possible  through  the  use  of  computers.  For 
the  federal^  airways  system  these  computers  would  operate  in  conjunction  with  the  VHF 
omnidirectional  range  and  distance-measuring  equipment.  With  the  computer,  it  is 


D  c  signals  from  locallzer 


Cross  pointer 
nstrument 

Vibrator 

400rul 
•  —  --'ill 

Jrr\ 

D-c  signals  from 

It-fir 

v  y- 
To  sim 
coupling 
for  thrc 
or  elev 

—  / 
ar 
unit 
ttle 
ator 

\1JJ 

DC      1 
DC  -nf 

Smoothing 

Rate  derivation 

Conversion  dc  to  ac 

—  *vWv>--f-  *•—  jfn 

Ac 

Conversion  of  a-c              p^H 
signal  amplitude                Fbeaecdk" 
Into  equivalent 
mechanical  rotation 

nc 

Ac 
Preamplifier 

"t- 
i 

If 

Servo 

amplifier 

Anti- 
hunt 

I        I 

Servo  motor 
&  gear  box 

Vetoc. 
gen. 

$ 

400  <\J 


Conversion  of 
mechanical 
rotation  Into 
auto-pilot  signal 


Auto-pilot 
system 


Bank  limit  adj. 


40000 


FIG.  22.     Auto-pilot  and  Radio  Coupling  Systems 

possible  to  fly  a  synthetic  left-right  indicated  course  in  any  direction,  whereas  the  regular 
courses  are  defined  only  toward  or  away  (radially)  from  the  station. 

Referring  to  Fig.  23,  S  represents  the  omnirange  station,  and  the  course  to  be  flown  lies 
along  the  non-radial  line  AB.  The  distance-measuring  equipment  (described  above) 
produces  a  voltage  ei  which  is  proportional  to  r,  the  distance  to  the  airplane.  This  is 
applied  to  a  selsyn  primary  (rotor)  which  is  connected  to  the  omnirange  azimuth  indicator. 
The  sinusoidal  output  of  the  selsyn  secondary  winding  is 

62  —  ei  sin  9 

because  it  is  designed  to  vary  sinusoidally  with  rotor  angular  displacement.  Since  ei  is 
proportional  to  r,  the  equation  may  be  written: 

62  =  &r  sin  6  =  ky 

The  voltage  eg  therefore  is  proportional  to  the  variation  in  y,  that  is  to  the  displacement 
of  the  airplane  from  the  line  AB.  This  displacement  voltage  may  be  balanced  by  a  fixed 
voltage  and  presented  on  the  usual  left-right  course  indicator  for  manual  flight,  or  it  may 
be  coupled  to  the  auto-pilot  for  automatic  flight  along  the  selected  line  AB. 


22-24 


AIDS  TO  NAVIGATION 


In  setting  the  computer  so  that  the  constant  k  will  be  properly  handled,  the  course  line 
to  be  flown  is  decided  upon  and  drawn  on  the  range  map.  The  direction  and  length 
of  the  perpendicular  from  the  range  station  to  this  line  are  determined.  The  direction  of 
the  perpendicular  is  set  into  a  calibrated  clutch  between  the  azimuth  indicator  and  the 
selsyn  unit.  The  length  of  the  perpendicular  line  is  set  up  on  the  "lane  selector"  switch, 
thereby  providing  the  required  fixed  balancing  voltage.  Then  the  pilot's  left-right  indica- 
tor will  be  centered  only  when  the  airplane  is  on  the  desired  flight  line. 


Azimuth  / 

clutch  v^,,/ 


Dtrectlor 

(0) 


indicator 


For  automatic 
flight  control 

FIG.  23.     Computer  for  Synthetic  (r0)  Courses 

RADAR  MONITOR  FOR  AIRPORT  TRAFFIC  CONTROL.  Civilian  aviation  will 
require  the  use  of  radar  as  a  monitor  in  airport  control  towers.  Development  of  suitable 
equipments  is  under  way.  Its  first  use  will  probably  be  in  expediting  the  outbound  traffic 
at  the  congested  terminals.  The  airport  tower  operator,  having  accurate  knowledge  of 
the  displacement  of  all  inbound  airplanes,  through  reference  to  the  radar  monitor,  may 
permit  the  departure  of  airplanes  that  otherwise  would  be  required  to  await  position  reports 
from  the  inbound  airplanes. 

Three  fundamental  problems  are  evident  in  the  application  of  radar  to  airport  control 
towers.  These  problems  are  typical  of  all  civil  radar  application.  They  were  overcome 
in  military  use  by  a  multiplicity  of  radar  equipments  and  large  operating  crews.  The 
problems  are:  (1)  reliable  coverage  up  to  about  30  miles,  from  horizon  to  zenith;  (2)  the 
elimination  of  undesirable  ground  clutter;  (3)  the  presentation  of  resulting  radar  informa- 
tion in  a  manner  such  that  the  regular  tower  crew  can  use  it  safely,  at  any  instant  day  or 
night,  without  special  enclosures. 

Relatively  high-angle  coverage  has  been  obtained  by  means  of  antennas  giving  cosecant- 
squared  patterns,  shown  in  Fig.  24.  When  the  energy  is  distributed  in  the  required  pat- 
tern, the  maximum  distance  range  is  naturally  reduced.  A  peak  power  output  of  about 
0.5  megawatt  is  required  to  give  the  30-mile  horizontal  coverage  with  a  cosecant-squared 
.pattern. 

.  Some  work  has  been  done  on  the  elimination  of  ground  clutter  as  illustrated  in  Fig.  25. 
Only  moving  targets  are  permitted  to  appear.  Development  is  being  continued  to  sim- 
plify the  equipment  and  reduce  the  maintenance  required  to  keep  it  in  perfect  adjustment. 


FACILITIES  IN  THE  NEW  FEDERAL  AIRWAYS  SYSTEM      22-25 

The  presentation  feature  has  not  yet  been  satisfactorily  solved.  New,  high-intensity 
cathode-ray  tubes  are  being  made  available  as  an  aid  to  the  daylight  presentation  prob- 
lem. Another  means  of  getting  the  radar  scope  picture  visible  in  daylight  is  comple- 


12       3       456       78      9     10     11    12    13     14    15    16    17    18    19    20 
Distance  In  miles 

FIG.  24.     Cosecant-squared  Pattern  of  GCA  Search.  Antenna 

mentary  light  niters.  For  example,  if  the  control  tower  window  glass  is  colored  blue  and 
an  amber  filter  is  used  over  the  face  of  the  radar  scope,  the  picture  will  appear  the  same 
as  at  night.  The  reflections  of  the  observer's  face  can.  be  eliminated  by  installing  the  amber 
filter  at  a  45°  angle. 


FIG  25     Radar  10-mile  PPI  Scope  Pictures,     (a)  Heavy  ground  clutter  from  objects  surrounding'the 
station  "at  the  Indianapolis  Airport.     The  targets  between  60°  and  100°  are  buildings  in  Indianapolis. 


22-26 


AIDS  TO  NAVIGATION 


FIG.  25. 


(6)  Same  as  (a)  except  that  Ground  Clutter  Is  Removed  and  Four  Airplanes  Are  Shown  in 
Plan  Position  (Courtesy  CAA) 


5.  PROPOSED  NEW  LANDING  SYSTEMS 

GROUND  CONTROLLED  APPROACH  (GCA).  At  the  beginning  of  the  war,  a 
talk-down  system  was  conceived  for  landing  airplanes  in  bad  weather.  It  became  known 
as  GCA  because  it  served  in  reverse  to  other  systems — it  controlled  from  the  ground. 
Several  sets  of  GCA  were  put  in  service  before  the  end  of  the  war.  Many  sensational 
landings  of  military  airplanes  were  made  with  GCA,  saving  lives  and  valuable  property. 
Its  use  required  no  new  airborne  equipment  but  depended  only  upon  communications  and 
a  cooperative  pilot. 

The  GCA  equipment  is  a  composite,  trailer-type  station  with  three  radar  systems  and 
complete  communications  equipment.  It  is  placed  about  500  ft  to  the  side  of  the  runway 
and  at  the  end  opposite  from  that  on  which  the  landing  is  to  be  made.  Air-conditioning 
and  power-generating  equipment  are  carried  on  the  towing  truck. 

Originally  the  GCA  trailer  required  a  crew  of  five  operators.  Four  of  them  constantly 
watched  four  radar  scopes,  and  the  fifth  served  as  final  approach  controller  to  give  heading 
and  rate-of-descent  instructions  to  the  pilot  on  the  final  approach.  Now,  however,  the 
operation  of  landing  a  single  airplane  may  be  handled  by  one  operator  who  shifts  his 
attention  from  search  to  precision  scopes  as  the  airplane  orients  and  approaches  the  run- 
way, Fig.  26.  If  more  than  one  airplane  is  involved,  more  operators  are  required. 

The  function  of  GCA,  whether  operated  by  one  or  more  men,  is  as  follows :  (a)  search  for 
aircraft  in  all  directions,  using  PPI  scope  presentation;  (fe)  direct  aircraft  into  the  landing 
sector  about  6  miles  from  the  runway  at  1500  ft  elevation,  on  the  basis  of  search  radar 


PROPOSED   NEW  LANDING   SYSTEMS 


22-27 


information;  (c)  land  the  airplane  by  giving  the  pilot  explicit  instructions  constantly 
during  the  approach  as  to  heading  and  rate  of  descent,  on  the  basis  of  precision  azimuth 
and  elevation  radar  information. 

The  search  function  in  GCA  is  obtained  at  10  cm  by  an  antenna  rotating  at  30  rpm.  The 
antenna  is  a  special  reflector  with  bvo-dipole  array  fed  by  rectangular  wave  guide.  Its 
radiation  pattern  is  illustrated  in  Fi£.  24. 

The  precision  radar  system  consists  of  one  transmitter  (3  cm)  sharing  time  with  two 
special  antennas.  One  antenna  is  vertical  and  the  other  horizontal.  Each  antenna  con- 
sists of  a  multiplicity  of  collinear  dipoles  mounted  along  a  wave-guide  section  and  fed  from 


3  Mile 
Precision  scopes 


10  Mile 
Precision  scopes 


r— 

-» 

j* 

"I 

J 

M 

Magnetron 
oscillator 

T-R 
box 

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Het. 
unit 

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Rec. 

Synch, 
unit 

Spare 

Spare 

Spare 

Spare 

I  Motor-dilvan  1 I 

1       beam  shift  device        | 

FIG.  26.     GCA  Search  and  Precision  Radar  Equipment  Arranged  for  One-man  Operation 

small  probes  projecting  into  the  wave  guide.  The  spacing  of  the  dipoles  and  phase  of 
their  currents  create  the  sharp  patterns  required  for  precise  direction  determination. 
These  patterns  are  aimed  along- the  approach  path  and  then  made  to  oscillate  rhythmically 
across  the  path  by  shifting  the  dipole  current  phases  (mechanically  distorting  the  rear  of 
the  wave  guide  through  a  motor-driven  mechanism). 

The  received  precision  radar  echoes  are  displayed  on  offset-center  PPI  scopes  as  illus- 
trated in  Fig.  26.  This  presentation  and  the  narrow  radiated  beams  permit  the  operator 
to  obtain  great  precision  in  the  observation  of  aircraft  displacement.  The  vertical  antenna 
gives  the  glide  path  displacement,  and  the  horizontal  antenna  gives  lateral  displacement. 

One  unique  feature  in  the  GCA,  which  is  used  in  certain  other  radar  equipments,  is  the 
optical  system  of  the  PPI  scopes.  The  operator  views  an  illuminated  map  through  a  45° 
glass  plate.  The  scope  is  directed  at  the  glass  from  a  position  complementary  to  the 
position  of 'the  map  below.  The  scope  appears  superimposed  on  the  map.  It  is  on  the 
map  that  the  correct  approach  line  is  inscribed  for  use  by  the  controller  in  detecting  devia- 
tion of  the  airplane.  .  .  . 

Three  important  features  of  GCA  are:  (1)  No  equipment  is  required  in  airplanes  for 
its  use  other  than  already  existing  communications.  All  aircraft  then  may  use  n.  (2)  Its 
precision  and  straightness  of  path  are  unaffected  by  conditions  surrounding  the  site,  such 
as  buildings,  hangars,  and  wires.  (3)  The  precision  is  not  entirely  in  the  system  but  partly 
in  the  ability  of  the  operator  to  bisect  the  target  image.  (4)  The  identity  of  aircraft  being 


22-28  AIDS  TO  NAVIGATION 

controlled  depends  largely  upon  the  skill  and  attention  of  the  operators.  (5)  Skill,  practice, 
and  experience  are  required  in  any  blind  approach  operation.  In  GCA  this  skill  is  on  the 
ground. 

NAVA GLIDE  (Federal).  The  navaglide  instrument  landing  system  being  developed 
by  the  Federal  Telecommunications  Laboratories  is  a  microwave  system  using  only  one 
frequency  channel.  The  four-directional  signals  of  the  glide  path  and  localizer  indications 
will  operate  on  this  frequency  channel  simultaneously  through  the  use  of  a  scheme  of 
subcarrier  modulation.  The  receiver  of  the  "Navar"  navigation  system  (described  below) 
may  be  used,  ultimately,  also  for  the  landing  signals.  Accurate  distance-measuring 
equipment  will  replace  the  markers  of  the  present  instrument  landing  system,  permitting 
use  of  automatic  landing  equipment. 

MICROWAVE  (Sperry).  A  complete  microwave  instrument  landing  system,  operating 
at  approximately  2600  Me,  has  been  developed  and  successfully  operated  experimentally 
for  several  years  by  the  Sperry  Gyroscope  Company.  This  development,  now  in  pro- 
duction status,  overcomes  objectionable  siting  and  wave-reflection  problems  existing  on 
systems  using  lower  frequencies.  Reasonably  small  radiating  systems  are  able  to  con- 
centrate and  confine  the  radiated  energy  to  the  approach  sector  and  avoid  nearby  buildings, 
wires,  or  hangars.  Reflections  from  these  objects  on  other  instrument  landing  systems 
causes  course  and  path  bends. 

The  present  success  of  the  system  may  be  attributed  to  the  development  of  the  klystron 
tube  (2K36/416)  for  airborne  receiver  use  and  to  the  development  of  crystal  control  for 
both  the  transmitting  and  receiving  equipment. 

The  complete  system  consists  of  two  ground  stations  (localizer  and  glide  path)  and  one 
airborne  receiver.  The  ground  stations  are  identical  in  electrical  circuits  and  have  equiva- 
lent outputs  of  about  70  watts  but  use  different  radiating  elements.  The  glide  path 
radiator  is  a  tilted  vertical  parabola  fed  by  two  wave  guides  through  a  mechanical  modu- 
lator. The  wave-guide  feeds  are  displaced  from  the  parabola  focus  in  opposite  directions 
and  therefore  produce  two  overlapping  patterns,  the  plane  of  overlap  being  inclined  up- 
ward at  the  desired  approach  glide  angle.  The  patterns  are  amplitude  modulated  600  and 
900  cycles  respectively.  The  parabola  is  narrow  to  give  wide  horizontal  coverage.  This 
permits  its  being  placed  safely  to  the  side  of  the  runway. 

The  localizer  transmitting  antenna  is  a  paraboloid,  giving  a  relatively  concentrated 
pattern  in  both  horizontal  and  vertical  planes.  It  is  equipped  with  a  central  vertical 
dividing  shield  to  separate  the  two  wave-guide  feeds.  The  radiated  patterns  overlap  in  a 
vertical  plane  thereby  forming  a  course  for  left-right  guidance  along  the  runway  center 
line.  The  localizer  is  placed  on  the  runway  center  line  at  a  safe  distance  from  the  end 
opposite  from  that  on  which  the  airplane  lands. 

The  one  receiver  used  in  the  airplane  has  a  common  crystal-controlled  klystron  tube 
oscillator  serving  two  separate  IF  circuits.  The  filtered  and  rectified  outputs  of  the  two 
channels  are  balanced  and  connected  to  the  localizer  and  glide  path  pointers  respectively 
of  the  conventional  crossed-pointer  landing  instrument.  Two  flag  alarms  on  the  instru- 
ment serve  as  indicators  to  warn  against  failure  of  the  localizer  or  glide  path  signals. 

An  electronic  coupling  device  now  in  production  is  used  to  couple  the  localizer  and  glide 
path  signals  into  the  Sperry  (A-12)  electric  auto-pilot  for  automatic  approach  flight. 
It  works  equally  well  on  the  CAA  instrument  landing  system. 

6.  PROPOSED  NEW  SHORT-RANGE  NAVIGATION  SYSTEMS 

LANAC  (Hazeltine).  The  word  "Lanac"  is  derived  from  *  laminar  navigation  and  anti- 
collision." 

This  system  is  proposed  for  the  1000-Mc  band  and  utilizes  in  each  airplane  an  interroga- 
tor and  a  replier.  The  replier  consists  of  a  pulse  receiver  and  transmitter  capable  of  auto- 
matically replying  to  interrogating  pulses  of  proper  frequency  and  coding.  The  code 
key  of  the  replier  varies  with  altitude  layers  in  a  prescribed  standard  manner,  through  the 
use  of  an  aneroid  cell. 

The  interrogator  includes  a  pulse  transmitter,  coded  to  challenge  the  repliers  of  other 
aircraft,  and  a  receiver  to  interpret  the  reply.  Normally  it  interrogates  on  the  code  equiva- 
lent to  its  own  altitude  so  as  to  provide  anticollision  and  traffic  control  safety.  By  means 
of  a  switch,  the  interrogation  code  can  be  varied  to  scan  the  traffic  in  altitude  layers  above 
or  below  the  level  being  flown.  The  interrogator  antenna  is  directional  and  rotatable  so 
that  direction  as  well  as  distance  and  altitude  of  replies  can  be  observed  on  the  L-type  (or 
possibly  PPI)  scope  used  in  the  system. 

Ground  transponders  and  ground  interrogator  stations  are  included  to  permit  route  navi- 
gation by  the  aircraft  and  ground  surveillance  of  airways  traffic  overhead,  obtaining  posi- 


PROPOSED  NEW  SHORT-RANGE  NAVIGATION  SYSTEMS      22-29 

tion  of  planes  in  three  dimensions  as  well  as  identity  of  each  plane,  and  affording  selective 
signaling  to  each  plane.  The  ground  transponders  also  serve  as  obstruction  warning  units. 
Ground  transponders,  properly  placed  on  the  approach,  aid  the  airplane  on  instrument 
approach  to  the  airport. 

The  Lanac  interrogator  may  be  employed  as  a  radar,  in  special  applications,  by  tuning 
the  receiver  and  transmitter  of  the  interrogator  to  the  same  frequency  so  that  echoes  will 
be  shown  on  the  interrogator's  display  in  the  usual  radar  manner.  In  this  case,  of  course, 
no  replier  is  utilized. 

This  radar  mode  of  operation  is  used  in  an  aircraft  to  supply  terrain-clearance  informa- 
tion when  the  plane  is  flying  at  an  altitude  of  500  ft  or  more.  It  therefore  offers  an  im- 
portant safety  feature  for  off-airway  flying  over  mountainous  country.  The  radar  mode 
is  used  in  marine  anticollision  service  to  warn  against  unequipped  craft.  Radar  anticol- 
lision  protection  between  aircraft  is  not  feasible  because  of  the  small  size  of  the  targets  and 
their  extremely  high  relative  speeds  when  closing  on  a  head-on-collision  course. 

The  Lanac  system  is  useful  in  marine  navigation,  and  a  description  of  this  application  is 
given  under  Marine  Aids. 

TELERAN  (RCA)  (ieZevision  radar  air  navigation) .  Teleran  is  a  comprehensive  system 
of  navigation  involving  radar  as  the  means  of  collecting  air  traffic  information  and  tele- 
vision as  a  means  of  displaying  the  information  to  the  pilot.  Radar  stations  with  over- 
lapping 50-mile  service  areas  to  form  the  airway  are  proposed.  Respondors  in  every 
airplane  are  essential,  although  failures  of  respondors  can  be  taken  care  of  in  the  plan  by 
using  separate  echo-type  search  radar  equipment.  The  PPI  picture  obtained  in  the  ground 
radar  is  used  by  air  traffic  control  or  tower  operators.  This  same  picture,  with  any  de- 
sired obstruction,  control,  or  weather  instructions  superimposed,  is  televised,  transmitted 
to,  and  repeated  in  the  airplane  by  television. 

The  system  permits  both  pilot  and  ground  operator  to  see  and  appreciate  the  complete 
air  traffic  situation.  The  airborne  transponder  units  can  be  coded  or  varied  automatically 
with  altitude  to  segregate  the  various  flight  levels  and  to  permit  identification  of  aircraft. 
Instructions  may  be  written  out  on  the  ground  and  "handed"  to  the  pilot  on  the  television 
picture. 

For  landing  at  an  airport,  an  airport  localizer  radar  and  GCA  precision  radar  units  are 
proposed. 

For  each  altitude  level  and  for  the  echo-type  search  radar,  separate  television  trans- 
mitting equipment  and  radio-frequency  channels  are  required.  The  channels  required 
can  be  greatly  reduced  over  commercial  television  because  a  low  scanning  rate  may 
be  used. 

NAVAR  (Federal)  (navigational  and  traffic  control  radar).  A  system  for  traffic  control 
and  navigation  around  airports  and  along  airways.  This  system  provides  the  following 
features,  which  may  be  applied  progressively  to  an  airways  system:  (a)  Ground  radar 
surveillance  in  the  form  of  PPI  displays,  (b)  Distance  and  azimuth  information  in  the 
airplane.  The  azimuth  information  is  omnidirectional  derived  from  the  ground  radar 
system,  (c)  PPI  traffic  presentation  in  the  airplane,  relayed  from  the  ground  station. 
(d)  Selective  signaling,  ground  to  aircraft;  and  automatic  identity  and  altitude  response, 
aircraft  to  ground. 

The  ground  radar  equipment  and  display  is  conventional  except  that  provision  is  made 
to  separate  known  (respondor-equipped)  aircraft  from,  others.  The  distance  information 
in  the  airplane  is  obtained  by  a  pulse  interrogating  system.  The  azimuth  information  in 
the  airplane  is  obtained  by  measuring  the  time  from  the  reception  of  a  non-directional 
pulse  radiated  from  the  ground  to  the  reception  of  the  rotating  search  radar  beam.  The 
pulse  and  beam  are  synchronized  for  true  north  direction.  The  timing  measurement  in 
the  airplane  is  made  automatically. 

The  airborne  PPI  display,  called  "Navascope,"  is  obtained  by  sending  synchronized 
pulses  omnidirectionally  from  the  ground  radar  station.  These  pulses  contain  the  in- 
formation of  all  aircraft  in  the  area  as  revealed  on  the  ground  radar  scope  and  are  repro- 
duced in  synchronism  on  the  airborne  PPI.  The  airborne  PPI  display  includes  positive 
identification  of  observer's  own  airplane  and  self-centered  altitude  layer  presentation. 

Selective  signaling  of  aircraft  is  obtained  by  directing  a  sharp  "challenging"  beam  in 
the  direction  of  the  airplane  on  which  a  check-up  is  desired.  The  aircraft  respondor 
beacon  circuit  is  to  contain  a  double-pulse  gate.  The  gate  is  tied  in  with  the  aircraft 
distance  indicator,  so  that  it  varies  with  distance.  The  interrogating  pulses,  beamed  from 
the  ground,  are  spaced  automatically  as  the  operator  selects  the  distance  range  of  the 
target  he  wishes  to  challenge.  Thus  the  challenge  is  narrowed  down  to  distance  and 
direction. 

The  complete  system  proposes  one  airborne  transmitter  and  two  receivers.  This  does 
not  include  communications  equipment. 


22-30  AIDS  TO  NAVIGATION 

MICROWAVE  OMNIDIRECTIONAL  RADIO  RANGE  (Sperry).  A  radio  range 
operating  on  the  same  general  principles  as  described  for  the  CAA  VHF  omnidirectional 
range,  but  in  the  microwave  frequencies  (2600  Me),  has  been  developed  by  the  Sperry 
Gyroscope  Company.  Like  the  CAA  system,  it  utilizes  the  comparison  of  phase  between 
two  audio  waves  in  the  airplane  to  determine  the  azimuth  bearing  to  the  station.  In 
the  microwave  system,  however,  the  antenna  of  the  ground  station  is  so  small  that  it 
can  be  rotated  at  1800  rpm,  thereby  avoiding  the  use  of  a  capacity  goniometer  and  avoid- 
ing antenna  phasing  problems. 

In  this  system,  as  in  the  Sperry  microwave  instrument  landing  system,  the  cw  signal 
to  be  radiated  is  generated  by  a  crystal-controlled  klystron  tube.  A  synchronous  1800-rpm 
motor  drives  a  small  alternator,  the  output  of  which  frequency-modulates  a  70-kc  sub- 
carrier.  The  subcarrier,  as  in  the  CAA  system,  amplitude-modulates  the  microwave 
carrier.  The  modulated  carrier  is  conducted  to  an  antenna  system  consisting  of  a  vertical 
stack  of  three  small  loop  antennas.  The  stack  gives  vertical  directivity  for  maximum 
horizontal  coverage.  The  modulated  carrier  radiation  produces  the  reference  audio 
voltage  in  the  airborne  receiver.  Its  relative  phase  is  the  same  in  all  azimuthal  directions. 

The  vertical  stack  of  antennas  also  has  a  directive  pattern  in  the  horizontal  plane.  This 
horizontal  pattern  is  essentially  sinusoidal.  By  rotating  the  pattern,  with  the  same  motor 
as  used  above,  another  audio  voltage  is  generated  in  the  receiver.  This  voltave  wave, 
with  respect  to  the  former  reference  voltage  wave,  has  a  phase  that  varies  with  position 
around  the  station.  For  the  full  360°  around  the  station  there  is  a  complete  cycle  or  360° 
phase  variation  in  the  wave.  This  is  called  the  variable  signal.  By  comparing  the  phase 
between  this  variable  and  the  former  reference  signal  the  bearing  from  the  station  can  be 
determined. 

The  phases  may  be  compared  in  the  same  manner  as  in  the  case  of  the  CAA  range,  or 
the  phase  detector  may  be  used  to  control  the  operation  of  a  motor  which  turns  a  map  in 
accordance  with  movement  of  the  airplane  around  the  station. 

The  proposal  for  the  complete  system  includes  the  addition  of  distance-measuring  signals 
and  communications  through  the  single  ground  station  transmitter  and  antenna. 

The  greatest  advantage  in  the  use  of  the  proposed  system  is  the  elimination  of  the 
goniometer  through  the  rotation  of  the  antenna  system.  Reduction  in  the  amplitude  of 
course  bends  due  to  siting  (reflections  from  buildings,  trees,  etc.)  are  evident. 

AEROTRONICS  (Raytheon).  The  Aerotronics  system  is  a  proposal  based  on  radar 
technique  and  uses  PPI  scope  presentation  in  the  airplane  and  on  the  ground.  It  includes 
(1)  airborne  radar,  (2)  ground  beacons  (transponders),  (3)  ground  radar,  (4)  airborne 
transponders,  and  (5)  distance-measuring  equipment.  With  this  equipment,  navigation, 
collision  prevention,  airways  traffic  control,  approach,  and  landing  are  to  be  taken  care  of 
without  other  aids. 

The  airborne  equipment  includes  a  rotating  antenna  for  radar  search,  an  omnidirec- 
tional antenna  for  communications,  and  an  antenna  for  distance  measuring.  The  ground 
radar  includes  both  azimuth  and  elevation  search.  This  is  for  traffic  surveillance  and 
control.  Ground  beacons  (transponders)  would  be  used  for  route  navigation. 

The  distance-measuring  equipment  proposed  operates  on  the  principle  of  the  optical 
interferometer.  It  operates  a  counter  at  the  ground  traffic  control  station  where  distance 
to  the  airplane  may  be  observed  with  great  accuracy. 

MULTIPLE  TRACK  RADAR  RANGE  (Australian).  A  multiple  track  radar  range 
(MTR)  has  been  developed  and  successfully  demonstrated  by  the  Council  for  Scientific 
and  Industrial  Research  of  Australia  at  Sydney.  This  range  produces  visually  indicated 
(left-right) ,  positively  identified  flight  tracks,  based  on  the  time  of  arrival  of  pulses  from 
two  spaced  ground  stations.  VHF  (212  Me)  is  employed,  which  results  in  line-of-sight 
distance  coverage.  The  tracks  are  hyperbolic  in  shape,  but  essentially  straight  beyond 
20  miles.  The  tracks  or  courses  are  fly  able  to  within  about  ±1°. 

The  MTR  system  is  fundamentally  the  same  as  GEE,  a  British  development  used 
extensively  in  the  invasion  of  Europe,  except  for  type  of  presentation.  GEE  is  a  hyperbolic 
system  similar  to  Lor  an.  except  for  its  frequency.  Two  stations  only  are  used  in  the  MTR 
system,  one  a  master,  sending  a  series  of  equally  spaced  pulses  (5000  pps  in  this  case)  of 
peak  power  about  10  kw.  The  second,  a  slave  station,  is  equal  to  the  master,  but  spaced 
about  8  miles  distant  and  normally  inoperative.  The  slave  station  receives  the  pulses 
from  the  master  and  rebroadcasts  them  with  a  suitable  fixed  time  delay.  For  any  position 
around  the  pair  of  stations,  the  pulses  are  received  in  a  specific  time  relationship.  The 
contours  of  equal  time  differences  are  hyperbolas  passing  between  the  stations  and  are 
used  as  tracks  or  courses.  Non-directional,  vertically  polarized  antennas  are  used. 

In  the  airborne  equipment,  the  pulse  signals  are  compared  in  an  automatic  circuit 
similar  basically  to  that  in  the  distance-measuring  equipment  described  before.  Standard 
time  differences  chosen  as  the  tracks  are  set  up  on  a  calibrated  switch  dial  and  numbered 


MISCELLANEOUS  RADIO  AIDS  22-31 

(track  numbers  1  to  30) .  When  a  given  track  is  selected,  a  left-right  indicator  signals  the 
pilot  any  deviation  from  that  track. 

The  master  and  slave  stations  are  separated  in  the  receiver  by  permitting  the  master 
(only)  to  transmit  double  pulses.  MTR  equipments  at  adjacent  places  are  identified  by  a 
selected  difference  in  repetition  rate. 

Distance-measuring  equipment,  of  the  interrogator-respondor  type,  is  proposed  for  use 
with  the  MTR  system. 

7.  PROPOSED  NEW  LONG-RANGE  NAVIGATION  SYSTEMS 

CAA  LOW-FREQUENCY  OMNIDIRECTIONAL  RANGE.  The  low-frequency  omni- 
directional range  operates  like  the  VHF  omnidirectional  range,  i.e.,  in  the  comparison  of 
the  phase  of  two  audio  signals.  In  early  tests  of  this  range,  two  individual,  basic,  carrier 
frequencies  were  used — one  at  172  and  one  at  194  kc. 

The  low-frequency  omnirange  consists  of  the  conventional  five-tower  Adcock  antenna 
array,  one  tower  at  each  corner  of  a  square  and  the  fifth  tower  in  the  center.  The  ref- 
erence signal,  172-kc  carrier,  modulated  with  30  cps,  is  fed  into  the  center  tower  only  and 
is,  therefore,  radiated  non-directionally.  The  other  signal  (194  kc)  is  fed  to  the  corner 
towers  through  an  inductive  goniometer  which  is  mechanically  rotated  at  1800  rpm.  The 
rotation  of  the  goniometer  spins  the  figure-of-eight  pattern  of  the  corner  towers,  producing 
the  variable  phase  signal  in  the  aircraft  receiver.  When  compared  to  the  phase  of  the 
reference  signal  in  a  wattmeter  circuit,  it  can  be  made  to  indicate  azimuth. 

Tests  are  being  conducted  using  a  single  low  frequency  and  a  subcarrier  of  1000  cps. 
The  30-cps  reference  is  applied  to  this  subcarrier  by  frequency  modulation,  as  in  the  case 
of  the  VHF. 

The  wattmeter  indicating  circuits  are  the  same  as  those  used  with  the  CAA  VHF  omni- 
receiver, 

Night  effect  represents  a  serious  problem  for  radio  ranges.  The  lower  the  frequency 
employed,  the  greater  is  the  ground  wave  signal  strength  and  the  more  stable  will  be  the 
operation  of  the  system.  Night  effect  is  minimized  by  curtailing  the  amount  of  unwanted 
vertical  radiation.  The  fact  that  the  signal  arrives  via  ground  or  sky  does  not,  in  itself, 
introduce  an  error  in  the  omni  system  since  the  indication  does  not  depend  upon  the  length 
of  the  path.  The  development  may  result  in  much  more  freedom  from  night-effect  errors 
and  swinging  than  in  an  automatic  direction  finding  system  employing  a  loop  and  operating 
on  the  same  frequency. 

SONNE  (Consul).  This  is  a  German  aural  CW  system  used  against  the  Allies  during 
the  war  and  capable  of  great  accuracy  over  long  distances  and  useful  in  both  air  and 
marine  service.  For  its  description,  refer  to  Marine  Aids,  article  10. 

LORAN  (Z0ng-range  air  navigation).  Excellent  results  are  being  obtained  in  the  ex- 
tensive use  of  Loran  in  transoceanic  flights.  For  a  description  of  this  system,  refer  to 
Marine  Aids,  article  10. 

NAVAGLOBE  (Federal).  This  is  a  CW  system  using  very  low  frequencies  for  long- 
range  navigation  over  oceans  and  continents.  The  receiver  proposed  has  a  narrow-band, 
noise-rejecting  feature  and  ADF  facilities.  The  ground  station  consists  of  three  antennas 
spaced  in  the  corners  of  an  equilateral  triangle,  and  a  transmitter  which  is  connected  in 
succession  to  the  three  antenna  pairs  each  second.  Three  dumbbell-shaped  patterns, 
displaced  in  bearing  by  120°  from  each  other,  are  radiated  successively.  Relative  ampli- 
tudes of  the  three  successive  signals  received  during  each  cycle  are  measured  automatically 
by  a  ratiometer,  permitting  direct  visual  indication  of  bearing  at  all  azimuths  from  the 
station.  For  other  directions  between  courses,  measurement  of  ratio  in  a  ratiometer  in 
the  receiver  permits  determination  of  relative  "bearing  with  respect  to  the  antenna  array. 

8.  MISCELLANEOUS  RADIO  AIDS 

VERTICAL  SEPARATION  INDICATOR.  The  Stratoscope  is  an  instrument  con- 
ceived to  provide  visual  indication  in  an  airplane  and  on  a  ground  monitor  of  the  vertical 
separation  between  airplanes  (or  other  obstacles)  within  a  minimum  service  area  of  10 
miles  distance  and  1000  ft  elevation. 

Basically,  the  Stratoscope  operates  by  converting  aircraft  height  into  frequency  and 
then  utilizing  panoramic  reception  to  display  received  signals  along  a  CRT  time  base 
calibrated  in  relative  vertical  height.  The  airplane  equipment  includes  a  transmitter 
and  a  receiver-indicator  unit.  The  ground  monitor  needs  only  a  receiver-indicator 
unit. 


22-32  AIDS  TO  NAVIGATION 

Conversion  of  height  into  frequency  is  accomplished  by  means  of  a  precision  aneroid 
cell  which  operates  a  variable  tuning  condenser  in  a  coaxial  line  oscillator.  The  oscillator 
frequency  changes  5  Me  in  10,000  ft.  This  change  is  accomplished  in  the  frequency  band 
of  148-154  Me.  The  height  of  the  transmitter  in  relative  pressure  altitude  may  therefore 
be  determined  by  measurement  of  transmitter  frequency. 

Interference  from  the  plane's  own  transmitter  is  avoided  by  sharing  time  between  the 
transmitter  and  receiver  at  an  alternation  rate  of  30  cps.  By  coding  the  transmission, 
some  identity  of  the  airplane  can  be  provided  in  the  receiving  indicators. 

In  tests  of  this  equipment  an  accuracy  of  200  ft  was  observed.  The  required  10-  or  20- 
mile  distance  range  can  be  covered  with  about  5  watts  (transmitter  power  output). 

ABSOLUTE  ALTIMETERS.  Absolute  altimeters  have  little  utility  in  navigation  over 
land  because  of  the  rapid  variation  in  indication  in  rough  territory,  but  they  do  have 
utility  in  navigation  over  water,  which  is  flat,  as  a  means  of  maintaining  constant  height 
above  the  surface.  Variations  then  in  the  pressure  altimeter  indicate  flight  toward  or 
away  from  atmospheric-pressure  areas.  Flights  directed  laterally  so  as  to  give  constant- 
pressure  altimeter  reading  (for  constant  absolute  altitude)  avoid  storm  areas  over  the 
oceans. 

Many  absolute  altimeters  have  been  invented  over  a  period  of  twenty  years,  including 
sonic,  capacity,  frequency  modulation,  and  radio-pulse  types,  A  reference  is  given  for 
detailed  information  as  space  permits  only  summary  remarks  here. 

One  absolute  altimeter,  called  a  "terrain  clearance  indicator,"  was  developed  for  com- 
mercial use  just  before  the  war.  It  operates  on  the  FM  principle  at  about  432  Me.  The 
transmitter  continuously  radiates  its  energy  downward  from  a  doublet  antenna.  The 
frequency  is  varied  from  410  to  445  Me  sixty  times  per  second.  The  receiver  takes  in  the 
energy  reflected  back  from  the  ground  and  also  some  of  the  original  transmitted  energy. 
The  receiver  output,  which  may  be  considered  a  beat  between  the  two  signals,  has  a 
frequency  depending  upon  and  increasing  with  altitude. 

The  FM  type  absolute  altimeter  is  capable  of  operating  down  essentially  to  zero  altitude. 
It  has  greatest  accuracy  at  lowest  altitude  but  is  subject  to  possible  serious  errors  from 
adjustment  and  noise  which  prevent  its  immediate  use  on  instrument  landing. 

The  radiopulse  altimeter  developed  during  the  war  for  high-altitude  bombing  uses  a 
cathode-ray  tube  as  a  means  of  displaying  time  between  transmitted  pulse  and  echo.  The 
display  is  calibrated  in  feet  of  altitude  and  arranged  circularly  around  the  tube  face. 
Essentially  it  is  a  radar  distance  indicator  with  its  antenna  fixed  in  a  position  under  the 
airplane.  The  pulse  type  altimeter  is  extremely  accurate  at  high  altitude  but,  at  present, 
useless  at  altitudes  below  several  hundred  feet.  Future  development  may  bring  about 
reduction  in  this  minimum  altitude. 

REFERENCES 

GOVERNMENT— CAA 

The  U.  S.  Dept.  of  Commerce — How  It  Serves  You  on  Land  and  Sea  and  in  the  Air.     U.  S.  Government 

Printing  Office  (January  1946). 

The  FCC  Allocation  Plan,  Electronics,  March  1945,  p.  92. 
Advancing  Air  Navigation.     U.  S.  Government  Printing  Office  (1946). 


FOREIGN 

xline  Navigation  Was  Different  in 

0 ,  Aircraft  Radio  Eauit>ment  for  Us 

p.  979. 


Gravis,  Airline  Navigation  Was  Different  in  Europe,  Aviation  Mag.,  November  1940,  p.  40. 
Hodgson,  Aircraft  Radio  Equipment  for  Use  on  European  Air  Lines,  Proc.  I.R.E.,  September  1935, 


RADIO  RANGES— LF 

Diamond,  On  the  Solution  of  Night  Effects  and  the  Radio  Range  Beacon  System,  Proc.  I.R.E.,  June 

1933,  p.  808. 
Jackson-Stuart,  Simultaneous  Radio  Range  and  Telephone  Transmission,  Proc.  I.R.E.,  March  1937, 

p.  314. 
Army-Navy  Precipitation  Static  Project,  Parts  I  to  VI,  Proc.  I.R.E.,  April  1946,  p.  156P;  May  1946, 

p.  234P. 

RADIO  RANGES— VHF 

An  Ultra-high-frequency  Radio  Range  with  Sector  Identification  and  Simultaneous  Voice,  Electrical 

Comm.,  June  1946,  and  Proc.  I.R.E.,  January  1946,  p.  9W. 
Development  of  the  VHF  Radio  Range,  Part  III,  U.  S.  Dept.  of  Comm.  CAA  T.  D.  Report  49. 

RADIO  MARKERS 

Jackson,  Metz*  McKeel,  Test  of  First  Manufactured  Fan  Marker,  U.  S.  Dept.  of  Comm.  CAA  T.  D. 

Report  15  (July  1938). 

Hromada,  Development  of  New  Station  Location  or  1  Marker,  Proc.  I.R.E.,  August  1944,  p.  63. ' 
LaPort,  Radiating  System  for  75  Me  Cone  of  Silence  Marker,  Proc.  I.R.E.,  January  1942,  p.  26., 


RADIO  AIDS  TO  MARINE  NAVIGATION  22-33 

AUTOMATIC  DIRECTION  FINDERS 

Roberts,  H.  W.,  Aircraft  Direction  Finders,  Chapter  8,  p.  190.     Morrow,  New  York. 

COMMUNICATIONS 

Ellihorn,  Anti-noise  Characteristics  of  Differential  Mies,  Proc.  I.R.E.,  February  1946,  p.  84P. 

Shawn,  Application  of  the  Throat  Mic,  Communications,  January  1943,  p.  11. 

Bennett  et  al.f  The  Design  of  Broad  Band  Aircraft  Antenna  Systems,  Proc.  I.E.E.,  October  1945,  p.  671. 

AIR  TRAFFIC  CONTROL  AND  WEATHER 

Air  Traffic  Rules,  CAA  Manual  60.    U.  S.  Government  Printing  Office  (October  1945). 

Gilbert,  Future  Air  Traffic  _  Control,  Flying  Mag.,  April  1946,  p.  60. 

Diamond,  Recent  Applications  of  Radio  to  Remote  Indication  of  Meteorological  Elements  Elec.  Eng., 

April  1941,  p.  163. 

Hauck,  Radiosonde  Telemetering  Systems,  Electronics,  May  1946,  p.  120. 
Is  the  Air  Full,  Harper's  (July  1946). 

INSTRUMENT  LANDING 

Instrument  Landing  of  Aircraft,  Elec.  Eng.,  December  1940,  p.  495. 

Metz,  The  CAA-RTCA  Instrument  Landing  System,  U.  S.  Dept  of  Comm.  CAA  T.  D.  Reports  35 

and  36  (October  1943). 
Caporale,  The  CAA  Instrument  Landing  System,  Electronics,  February  1945,  p.  116;  March  1945, 

p.  128. 

Metz,  Radio  Glide  Path  for  Aircraft,  Radio  News,  November  1945,  p.  8. 
Montgomery,  A  VHF  Aircraft  Antenna  for  Reception  of  109-Mc  Localizer  Signals,  Proc.  I.R.E., 

November  1945,  p.  767. 

OMNIDIRECTIONAL  RADIO  RANGE 

Luck,  Omnidirectional  Radio-Range  System,  R.C.A.  Rev.,  March  1946,  p.  94, 
Stuart,  The  Omni-Directional  Range,  Aero  Digest  (June  15,  1945). 

RADAR 

The  Radar  Equation,  Electronics,  April  1945,  p.  92. 
Schneider,  Radar,  Proc.  I.R.E.,  August  1946,  p.  528. 

NEW  LANDING  SYSTEMS 

Ground  Controlled  Approach  for  Commercial  Aviation,  Electronics,  May  1946,  p.  160. 

Spicer,  The  GCA  Landing  Systems,  Bendix  Radio  Engineer,  January  1946,  p.  17. 

Folland,  The  Use  of  Microwave  for  Instrument  Landing,  Radior  April  1946,  p.  23. 

Aerial  Navigation  and  Traffic  Control  with  Navaglide,  etc.,  Electrical  Communication,  June  1946,  p.  113. 

ROUTE  NAVIGATION 

Herbst  et  al.,  The  Teleran  Proposal,  Electronics,  February  1946,  p.  124. 

Aerial  Navigation  and  Traffic  Control  with  Navar  and  Navaglobe,  Electrical  Communication,  June 
1946,  p.  113. 

Pierce,  Introduction  to  Loran,  Proc.  I.R.E.,  May  1946,  p.  216. 

Sandretto,  Absolute  Altimeters,  Proc.  I.R.E.,  May  1944,  p.  167. 

Frequency,  Power  and  Modulation  for  Long-range  Radio  Navigation  System,  Electrical  Communica- 
tion, June  1946,  p.  144. 


RADIO  AIDS  TO  MARINE  NAVIGATION 

By  M.  K.  Goldstein 

Table  1  shows  an  extensive  summary  and  classification  of  established,  recently  intro- 
duced, contemplated,  and  proposed  electronic  aids  to  marine  navigation.  It  will  be 
noticed  that  the  systems  are  classified  in  groups  dealing  with  radio  types,  sonic  and  super- 
sonic types,  and  red  and  infrared  types.  Also  the  various  systems  are  analyzed  for  essen- 
tial characteristics  falling  into  the  following  grouped  categories:  Type  of  system  (including 
basic  principle  and  position  information  supplied) ;  performance  of  system  (including  range, 
accuracy,  ambiguities,  operating  frequency,  and  band  width;  system  requirements  (in- 
cluding required  ship  and  shore  equipments) ;  and  the  system  usability  (including  engineer- 
ing status  and  reliability  factors) .  Details  concerning  each  of  the  specific  systems  classified 
are  discussed  in  the  following  articles  in  this  section. 


22-34 


AIDS  TO  NAVIGATION 


Table  1.    Summary  and  Classification  of  Established,  Recently 


System 

Type 

Performance 

Principle 
Employed 

Basic 
Information 
Supplied 

Maximum  * 
Range 

Accuracy  2 

Ambigu- 
ities 

Frequency 
(Mc/s) 

RADIO 

a.  Radiobeacon 

"Azimuth 

Short,  med. 

<2° 

None 

0.285-0.3153 

(shore) 
b,  Radiobeacon 
(ocean  sta.)  t 
c.  Radar  marker 
(Rarnark) 
d.  Respondor 

Non-directional 
transmission  and 
directional 
reception 

Azimuth 
•  Azimuth 
R  &  az.  fix 

Medium 
Optical 
Optical 

Devel'pm't'l 

±2%,  ±3° 

None 
None 

None 

0.30-0.55 
3,  5,  &  10  cm. 
3,  5,  &  10  cm. 

(Racon) 
e.  Reflector  (Racon)  ., 
f.  Rotating  beacon 

Dir.  trans.  &  non- 

R  &  az.  fix 
Azimuth 

Optical 
Medium 

±2%,  ±2° 

None 
Multiple  * 

3,  5,  &  10  cm. 
0.20-0.5 

(Consol.) 

dir.  reception 

2.  Direction  finding—  ship 

(Loop): 
a.  MF—  Aural  null    " 

"Azimuth     " 

X2° 

None 

0.3-3.0 

(Standard)  t 
b.  MF-Nuli  seeking 

Azimuth 

<2° 

None 

0.2-1.75 

(ADF) 

(SCR-269)  f 
c.  MF-Instantaneous 
(DAK)  f 
d.  MF—  Position  plot 

Directional 
reception  of  any 
transmission 

Azimuth 
Az.fix 

Normal 
communi- 
cation 
range 

'<2° 

None 
None 

0.25-1,5 
0.1-1.6 

(Bendix) 

e.  HF—  Instantane- 

Azimuth 

5°-10° 

None 

1.5-21 

ous  (DAQ, 

,  DAU)f 

. 

3.  Direction  finding—  shore:  t 
a.  MF-Adcock        i 

r  Azimuth     1 

M°-2° 

None 

0.25-1.5 

(DAH) 

Normal 

I 

b.  HF-Adcock 

Azimuth      ! 

communi- 

None 

1.5-30 

(DAJ.SCR-291) 
c.  HF  —  spaced   loop 
(DAB) 
d.  VHF-Adcock 

Directional 
reception  of  any 
transmission 

Azimuth     J 
Azimuth 

cation 
range 

Optical 

3°-4° 
3°-5° 

None 
None 

2.0-18 
100-160 

(DBF) 

e.  UHF—  Reflector 

Azimuth 

Optical 

3°-5° 

None 

90-5000 

(DBM)             J 

. 

4.  Radar: 

a.  Ship  

Echo  ranging 

R  &  az.  fix 

Optical 

±2%,  ±2° 

None 

3,  5,  &  10  cm. 

b.  Shore  

Echo  ranging 

Traffic 

Optical 

±2%,  ±2° 

None 

3,  5,  &  10  cm. 

5.  Propagationtimedifference 

a.  Loran  (stanoard)  .  " 

•HypLofP  7 

Long 

0.5% 

None8 

1.7-2.0 

b.  Loran  (SS)  10  

PTD  6—  synch. 

HypLofP 

Long 

0.5%-1.0% 

None?'  is 

1.7-2.0 

c.  Loran  (LF)11.... 

pulsed  trans- 
missions 

'  HypLofP 

Long 

<i.o% 

NoneS 

0.18 

d.  Gee  

Hyp  fix 

Optical 

<0.8% 

None  8 

20-85 

e.  Decca  j 

PPD  "-Synch. 

[Hyp  fix 

Medium 

±0.02%  i* 

Multiple" 

O.Ot-0.2 

f.  POPI  f 

CW  transmis- 
sions 

HypLofP 

Long 

Devel'pm't'l 

None 

0.75 

6.  Interrogator-respondor: 
a.  Shoran      .  .  . 

Dual  beacon 

Range  fix 

Optical 

±50  ft 

Few16 

210-320 

b.  Lanac—  ship  

ranging 
Single  beacon 

R&az.fix 

Optical 

±2%,  ±2° 

None 

1000 

ranging 

c.  Lanac  —  shore 

Single  beacon 

Traffic 

Optical 

±2%,  ±2° 

None 

1000 

ranging 

7.  Composite  data  relay: 

a  Teleran 

Instant,  data  relay 

Gen.  nav. 

Optical 

Telv.  Ch'n'l 

b.  Facsimile  

Recording  data 

Gen.  nav. 

Medium 

Facs'm'le 

relay 

Ch'n'l 

RADIO  AIDS  TO  MARINE  NAVIGATION 


22-35 


Introduced,  Contemplated,  and  Proposed  Aids  to  Marine  Navigation 


Performance 

Requirements 

Appraisal  Factors  * 

Signal 
Band  Width 

Minimum  Equipment  for  Basic 
Information  Supplied 

Engineering 
Status 

General  Remarks 

Ship 

Shore 

>CW 

DF  receiver 

Coded  beacon 

Proven 

Inaccurate  and  unreliable  beyond  predom- 

£CW 

DF  receiver 

Coded  beacon 

Proven 

inant  ground  wave  range. 
Combination    radiobeacon,     rescue     ship, 

weather  station. 

3-5  Me 

Special  radar 

Beacon 

Under  develop- 

Identified by  reference  to  natural  target 

ment 

3-5  Me 

Special  radar 

Respondor  beacon 

Proven 

Identification  obtained  by  coded  transmis- 

3-5 Me 
<lkc 

Radar  system 
Receiver 

Reflector 
1  station 

Proven 
Under  trials  6 

sion. 
Special  target  for  increased  echo  reflection. 
Minimum  range  —  25  mi;  accuracy  deteri- 

orates   at    night    beyond    predominant 

ground  wave  range. 

£CW 

Standard  DF              "| 

"Proven 

£CW 

Special  DF 

Proven 

2a  to  2e  inclusive—  unreliable  beyond  pre- 

:>CW 

Special  DF 
2  ADF'S  &  computer 

CW  or  other 
transmitter 

Proven 

Under  de- 
velopment 

dominant  ground  wave  range  due  to  sus- 
ceptibility to  polarization  error. 
Note  to  2d:  Fix  obtained  by  simultaneous 
cross-bearing  plot  on  map. 

£CW 

Special  DF 

Proven 

I 

£CW 

MF  transmitter 

Special  DF 

Proven 

£CW 

HF  transmitter 

Special  DF 

Proven 

3a  to  3e  inclusive  —  designated  models  have 

:>CW 

HF  transmitter 

Special  DF 

Proven 

low  susceptibility  to  polarization   errors 
beyond  ground  wave  ranges.    Indicated 

>cw 

VHP  transmitter 

Special  DF 

Proven 

accuracies  generally  realized  up  to  maxi- 
mum sky  wave  range. 

;>cw 

UHF  transmitter 

Special  DF 

Proven 

3-5  Mo 

Radar  system 

None 

Proven 

Valuable    anti-collision    and    short-range 

* 

naval  device. 

3-5  Ms 

Comm.  receiver 

Radar,  comm. 

Under  trials 

Suitable  for  traffic  control  in  congested  areas. 

transmitter 

50-70  kc9    - 

"2  stations 

Proven 

Day  range,  750  mi;  night  range,  1400  mi 

(over  sea  water). 

50-70  kc  9 

Special  receiver  and  indi- 

2 stations 

Under  trials 

Night  use  only;  utilizes  base  line  of  approx. 
100  mi. 

10  kc 

cator 

2  stations 

Under  trials 

Greater  ground  wave  range  and  more  stable 

1  Me  12 

-  3  stations 

Proven  5 

propagation  over  5a. 
Simultaneous  fix  from  2  hyperbolic  lines  ot 

position. 

3CWfreq's. 

Special  receiver 

3  stations 

Under  trials  20 

Limited  as  1  (a)  above  owing  to  ionospheric 

phase  shifts. 

<1kc 

Rcvr.  &  Spc'l  ind. 

1  station 

Under  develop- 
ment5 

Same  as  5e;  awaits  satisfactory  phase  meas- 
urement   between    sequentially    received 
signals. 

3-5  Me  v       } 

f  2  respondor 

Proven  u 

Bombing  and  mapping  aid;  may  have  marine 

1 
3-5  Me        [ 

Special  trans.  &  receiver 

J     beacons 
|  Respondor 

Under  trials 

use. 
Optional  operation:  radar  beacon  or  radar 

J 

[    beacon 

system. 

3-5  Me 

Respondor  beacon, 

Spc'l.  trans.  & 

Under  trials 

Traffic   control  utilizing  shipboard  trans- 

comm. receiver 

rcvr.,  comm. 

ponders. 

trans. 

3-5  Me 

Television  receiver 

Television 

Under  develop- 

Air traffic  control  system;  also  feasible  for 

transmitter 

ment 

marine  use. 

1-1  Okc 

Facsimile  recejver 

Facsimile 

Proven 

Permanent  recording  of  any  general  infor- 

transmitter 

mation,  data,  etc. 

22-36 


AIDS  TO  NAVIGATION 


Table  1.     Summary  and  Classification  of  Established,  Recently  Introduced, 


System 

Type 

Performance 

Principle 
Employed 

Basic 
Information 
Supplied 

Maximum  1 
Range 

Accuracy  2 

Ambigu- 
ities 

Frequency 
(Kc/s) 

SONIC  AND 
SUPERSONIC 
8.  Underwater  beacon  

Non-dir.  trans.  & 
dir.  reception 
Dir.  reception  of 
any  transmission 

Gen.  echo  ranging 
Vert,  echo  ranging 

PTD  6—  Explosion 
&  multiple  timed 
reception 

Non-dir.  trans.  & 
dir.  reception 
Dir.  reception  of 
any  transmission 
Optical  echo 
ranging 

Azimuth 
Azimuth 

R  &  az.  fix 
Depth 
Position  fix 

Azimuth 
R  &  az  fix 

V  short 
V  short 

V  short 
V  short 
Vlong 

V  short 
V  short 
U  short 

<r 

±1° 

<1%,  <1° 
<1% 
<5mi 

<1° 

<r 

Proposed 

None 
None 

None 
None 
None 

None 
None 
None 

0.8-30 
0.8-30 

15-30 
0.8-30 
0.05-1.0 

9.  Direction  finding 

10.  Sonar: 
a.  General  echo  ranging 

b.  Echo  sounding  
1  1    Sofar  t 

RED  AND 
INFRARED 
1  2   Beacon 

Microns  19 

Far  Infrared 
(0.7-13) 
Far  Infrared 
(0.7-13) 
Near  Infrared 
(0.3-1.5) 

1  3.  Direction  finding  
14.  Rcdar    ... 

*  Based  on  data  available  up  to  late  1949. 

t  Used  entirely  or  in  part  for  distress  service. 

1  Ultra-short 0-1  nautical  miles 

Very  short 1-10  nautical  miles 

Short 10-150  nautical  miles 

Medium 150-500  nautical  miles 

Long 500-1500  nautical  miles 

Very  long over  1500  nautical  miles 


2  Representative  accuracies  obtained  in  practice. 

3  Frequencies  in' current  U.  S.  use. 

4  Resolved  by  dead  reckoning  or  DF. 
6  British  only. 

6  Propagation  time  difference. 

7  Hyperbolic  line  of  position. 

8  Two  easily  resolved  ambiguities  can  exist  in  fix  obtained. 

9  Provisions  are  made  for  16  different  channels  on  the  same 
carrier  frequency  by  utilizing  different  pulse  repetition  rates. 


9.  ESTABLISHED  NAVIGATIONAL  AIDS 

RADIOBEACON  SYSTEMS  (MF)  (See  references  2  and  3.)  The  non-directional 
medium-frequency  (MF)  radiobeacon  system,  maintained  by  the  Coast  Guard  in  the 
United  States  and  by  similar  organizations  in  foreign  countries  throughout  the  world,  is 
the  most  extensively  used  radio  navigational  aid  today.  The  system  consists  of  fixed 
radiobeacon  stations  located  at  lighthouses,  lightships,  and  other  points,  which  transmit 
distinctive  identifying  coded  tone  signals,  enabling  navigators  at  sea  to  take  bearings  on 
them  by  means  of  medium-frequency  loop  direction  finders.  Bearings  obtained  from  two 
or  more  such  beacon  stations,  in  conjunction  with  charts  of  the  geographical  locations  of 
the  radiobeacons,  uniquely  establish  the  ship's  position.  A  single  radio  direction-finding 
(DF)  bearing  crossed  with  a  line  of  position  of  a  heavenly  body,  two  bearings  on  the  same 
station  and  the  distance  run  between  bearings,  or  a  bearing  and  a  synchronized  air  or 
submarine  fog  signal  also  suffice  to  determine  the  position  of  the  vessel.  In  the  last  case, 
blasts  from  the  sound  signal  are  synchronized  with  the  radiobeacon  signals  and  the  differ- 
ence in  time  of  reception  of  these  two  signals  can  be  converted  into  an  approximate  station 
distance.  Owing  to  the  many  factors  which  enter  into  the  transmission  and  reception  of 
radio  signals,  a  ship  cannot  estimate  its  distance  accurately  from  a  radiobeacon  either  by 
the  strength  of  the  signal  received  or  by  the  time  at  which  the  signals  were  first  heard. 
However,  a  line  of  direction  obtained  from  a  single  radio  bearing  enables  a  ship  to  proceed 
toward  the  radio  station  by  the  shortest  course.  This  is  especially  applicable  to  a  rescue 
ship,  enabling  it  to  head  directly  toward  the  ship  in  distress  and  thereby  arrive  in  the 
minimum  of  time.  Such  radio  bearings  are  usually  accurate  to  within  2°  or  less,  depending 
upon  the  equipment  and  the  operator's  skill.  DF  calibration  transmissions  are  available 
from  radiobeacons  upon  request,  continuous  signals  being  transmitted  while  the  ship's 
direction  finder  is  being  calibrated.  The  simplicity  and  reliability  .of  the  radiobeacon 
indicate  that  such  systems  will  continue  to  exist  as  a  navigational  aid  for  many  years. 

The  present  MF  radiobeacon  system  in  the  United  States  (see  Fig.  1  for  East  Coast 
and  Gulf)  consists  of  approximately  200  radiobeacons,  each  radiobeacon  operating  on 
one  of  the  even  frequencies  in  the  285-315  kc/s  band.  The  European  system  frequency 


ESTABLISHED  NAVIGATIONAL  AIDS 


22-37 


Contemplated,  and  Proposed  Aids  to  Marine  Navigation — Continued 


Performance 

Requirements 

Appraisal  Factors  * 

Signal 
Band  Width 

Minimum  Equipment  for  Basic 
Information  Supplied 

Engineering 
Status 

General  Remarks 

Ship 

Shore 

£CW 
£CW 

1-4  kc         I 
1-4  kc 
<  1000  eye 

Sonic  DF 
Sonic  DF 

Special  transmitter  and 
receiver 

Special  explosive  charge 

Infrared  DF 
Infrared  DF 

Special  transmission  and 
receiver 

Sonic  beacon 

Any  sonic 
transmission 

fNone 
JNone 

Special  receiver 
stations 

Infrared  beacon 

Any  infrared 
transmission 
None 

Proven 
Proven 

Proven 
Proven 
Tinder  trials 

Proven 
Proven 
Proposed 

Usefulness  limited  by  certain  underwater 
phenomena;  see  10  (a). 
Bearing  on  any  received  underwater  sound 
transmission;  see  8. 

Accuracy  subject  to  underwater  refraction 
and  propagation. 
Depth  finding  by  reflected  sound  transmis- 
sions. 
Proposed  distress  aid  using  natural  under- 
water sound  channel. 

Same  range  as  visual  light  under  fog  condi- 
tions. 
Same  range  as  visual  light  under  fog  condi- 
tions. 
Same  range  as  visual  light  under  fog  condi- 
tions. 

Microns 

Apr>rox.  13 
Approx.  13 
3.2 

10  Intended  to  increase  nighttime  accuracy  over  Standard 
Loran;  however,  comparable  accuracy  has  not  yet  been  prac- 
tically achieved. 

11  Intended  for  increased  daytime  range  over  Standard 
Loran;  however,  comparable  accuracy  has  not  yet  been  fully 
realized. 

12  If  necessary,  different  pulse  repetition  rates  will  permit 
multistation  operation  on  same  carrier  frequency, 

13  Propagation  phase  difference. 

14  For  medium  range. 


15  Resolution  awaits  satisfactory  system  of  "lane"  identi- 
fication. 

16  Solved  by  general  knowledge  of  position. 

17  Air  only. 

18  Four-station  operation  also  provided  to  give  increased 
accuracy  and  to  remove  ambiguities. 

19  The  thousandth  part  of  1  mm. 

20  A  British  Isle  Deeca  facility  is  in  24-hr  operation  with 
performance  results  indicating  a  proven  system. 


band  is  290-320  kc/s.  Distinctive  signals,  by  which  the  different  stations  are  positively 
identified,  are  obtained  by  keying  a  tone-modulated  carrier  to  give  simple  dot-dash  com- 
binations at  the  rate  of  thirty  characters  per  minute.  Because  of  the  great  number  of 
stations  and  the  restricted  frequency  band,  it  has  been  found  necessary  to  share  time 
between  stations  on  the  same  radio  frequency  by  transmitting  signals  from  any  one  radio- 
beacon  for  1  minute  followed  by  2  minutes  of  silence.  This  allows  two  adjacent  stations 
to  transmit  similar  signals  on  the  same  frequency  without  causing  interference  simply 
by  timing  the  transmissions  to  occur  during  successive  minutes.  Two-tone  modulation  is 
used  on  some  stations  in  congested  traffic  areas  to  add  distinctiveness  to  the  identifying 
signal.  During  clear  weather  the  radiobeacons  follow  such  cycles  of  transmission  for  one 
or  two  10-minute  periods  of  the  hour.  All  six  10-minute  periods  are  utilized  during  condi- 
tions of  fog.  The  beacons  are  carefully  timed  and  remotely  monitored. 

Radiobeacons  are  grouped  into  four  classifications,  according  to  their  power,  output  and 
maximum  effective  ranges: 

Class  A 750  watts— 200  miles 

Class  B 150  watts— 100  miles 

Class  C 25  watts— 20  miles 

Class  D 5  watts —  10  miles 

A  few  special  high-power  beacons  having  ranges  of  400  miles  have  also  been  put  into  opera- 
tion. Class  A  and  B  stations  require  identical  equipment  with  the  exception  of  the  750- 
watt  power  amplifier  used  with  the  A  stations.  The  transmitter,  which  is  coupled  directly 
to  the  antenna  for  the  B  station,  is  used  as  an  exciter  for  the  power  amplifier  of  the  A  sta- 
tion. Both  types  of  radiobeacons  often  use  125-ft  insulated  self-supporting  towers  as 
non-directional  antennas,  with  provision  for  location  up  to  several  hundred  feet  from  the 
transmitter  house  when  local  conditions  make  such  an  installation  preferable.  Sometimes 
an  insulated  guyed  antenna,  approximately  40  ft  high,  is  placed  on  top  of  a  light  tower 
above  the  lantern.  Lightships  use  vertical  or  symmetrical  T  antennas.  Inverted  L-type 
antennas  are  not  used  because  of  the  undesired  directivity  and  the  horizontally  polarized 
field  components.  The  frequency  of  the  A  and  B  stations  can  be  quickly  changed  to  any 


22-38 


AIDS  TO  NAVIGATION 


one  of  four  pre-set  frequencies  which  are  usually  crystal-controlled.  Class  C  radiobeacons 
are  used  for  short-range  navigation.  The  equipment  is  very  similar  to  that  of  a  class  A 
station  except  for  the  much  lower  power  and  the  reduction  in  size  of  equipment.  Owing 


For  CANADIAN  RADIOBFACONS 
7°"  «e  CANADIAN  LISTS.  «1 


<>        i 

MANANA  ID.  300  [Ml 

HALFWAY  ROCK  312  [3-D    01  SEAL  10   (CAN  I  301  [2-5] 


RACHOBEAC0N  SYSTEM     wate 

COAST 

'  SOSTON  Q1ST 

vv>  Off HJP  i  fitf&TQWHCWSIfipttlSN 

-  NEV\r>YQ&K  DISt 


MAKTUCKETL!  3M  fJ-J) (WARNING  BEACON) 


T ,  /^        A 

"j       I    A.  I F1YWB  PAN  LS,  2S*  [j.JJ 


UNITED  STATES  and  CANADIAN  radlqbeacons  are  assigned  group  I 
quencies  and  definite  operating  minutes.  The  sequence  within  a  group 
indicated  by  a  Roman  numeral  before  the  name,  thus  IT.  Stations  with  t 
same  sequence  numerals  transmit  on  the  same  minute. 


diobeacons  operate  during  fog  or  low  visibility  and  during  one  or  two 
10  minute  periods  out  of  each  hour  in  clear  weather,  (except  stations  at 
CANAL  APPROACHES  which  operate  on  scheduled  periods  only  and 
Cape  Henry  which  operates  continuously  In  assigned  sequence,  1  minute  on, 
2  minutes  off.  24  hours  daily. 


The  scheduled  10  minute  periods  of  each  hour  of  clear  weather  operation 
are  given  in  brackets  aftar  the  name  of  the  sution,  thus  [1-4]  [2-5]  etc 
(See  clock  face  diagram).  The  last  minute  of  each  clear  weather  10  minute 
period  Is  silent 


Washington.  D.C.,Mar.  1, 1946 


FIG.  1.     Typical  Radiobeacon.  Navigational  Chart,  Published  by  U.  S.  Coast  Guard 

to  the  necessity  of  making  the  signals  from  these  short-range  stations  especially  distinctive 
in  areas  of  considerable  marine  traffic,  the  modulating  audio  frequency  is  variable  in  four 
steps.  The  timer  cams  can  be  set  to  give  a  single  tone  or  any  combination  of  tones.  Class 
D  marker  radiobeacons  are  small  automatic  battery-operated  transmitters  located  on 
pierheads,  buoys,  ends  of  jetties,  etc.,  which  serve  as  local  markers  to  indicate  channel 
entrances,  turning  points,  etc.,  where  careful  approach  is  required.  They  are  not  syn- 


ESTABLISHED  NAVIGATIONAL  AIDS  22-39 

chronized  with  other  radiobeacons  but  operate  continuously  in  all  types  of  weather, 
sending  out  a  characteristic  of  several  dashes  during  each  30-sec  period.  The  transmitter 
frequency  is  crystal-controlled  and  has  a  modulation  frequency  of  1000  cycles  per  second. 
A  15-ft  welded  Monel  tripod  mast  is  normally  used  as  the  antenna.  Special  warning 
transmitters  are  located  at  some  radiobeacons;  each  transmitter  operates  on  the  same 
frequency  as  the  beacon  and  gives  a  distinctive  signal  to  warn  a  navigator  who  is  homing 
into  the  station  that  he  is  approaching  dangerously  close.  The  usable  range  of  these 
warning  beacons  is,  therefore,  intentionally  reduced  to  provide  warning  over  only  a  small 
but  sufficient  area  surrounding  the  long-range  beacon. 

At  present  the  characteristics  of  some  of  these  stations  are  being  changed  to  provide 
satisfactory  operation  with  automatic  direction  finders.  Instead  of  both  the  carrier  and 
modulation  being  interrupted  during  the  keying  process,  only  the  modulation  will  be 
interrupted.  In  addition,  many  of  the  stations  will  transmit  throughout  the  hour  under 
all  weather  conditions  rather  than  just  during  one  or  two  10-niinute  periods. 

The  well-established  aerobeacon  system  used  for  air  navigation  is  also  used  for  marine 
purposes  and  is  discussed  in  the  first  part  of  this  section. 

DIRECTION-FINDING  SYSTEMS.  (See  references  4,  5,  and  6.)  The  use  of  shipboard 
medium-frequency  radio  direction  finding  has  continued  to  grow  as  a  navigation  aid  since 
the  early  days  of  radio  broadcasting.  By  use  of  the  shipboard  direction  finder,  practically 
every  transmitting  station  within  the  frequency  band  and  range  of  the  radio  direction 
finder  is  a  potential  point  for  a  navigational  reference  line  of  position.  Shipboard  direction 
finding  also  plays  an  important  role  in  directly  guiding  vessels  to  other  vessels  in  distress. 
As  an  additional  aid  to  navigation  in  time  of  distress,  medium-frequency  direction-finding 
stations  Ipcated  along  the  coasts  and  on  the  Great  Lakes  are  available  to  mariners  who 
transmit  signals  on  the  international  distress  frequency  of  500  kc/s.  Various  other  fre- 
quencies are  available  for  distress  and  emergency  purposes,  depending  on  the  range  from 
land  or  the  location  of  the  vessel.  For  example,  8280  kc/s  is  used  for  United  States  long- 
range  contact  and  2182  kc/s  is  used  on  the  Great  Lakes. 

While  the  most  widely  used  shipboard  direction  finder  is  of  the  loop  aural-null  type, 
other  types  are  available  (see  Section  6,  article  32). 

Environmental  Effects.  (See  reference  5.)  The  term  "environmental  effects"  is  used  to 
describe  all  local  physical  conditions  which  cause  DF  bearing  errors  aboard  ship  and  at 
shore  stations.  Such  conditions  normally  involve  metallic  structures  near  the  DF  antenna 
and  their  radiation  fields,  induction  fields,  or  shielding  effects.  For  frequencies  below  1000 
kc/s,  shipboard  environmental  effects  can  be  sufficiently  well  controlled  or  compensated 
(further  details  are  given  by  C.  T.  Solt,  Proc.  I.R.E.,  Vol.  20,  p.  228,  February  1932)  so 
that  direction  finders  in  this  band  give  highly  satisfactory  performance  on  ground-wave 
transmissions;  i.e.,  the  resultant  calibration  curve  is  fairly  symmetrical  and  does  not 
exceed  a  few  degrees  maximum  deviation.  Arriving  signals  of  certain  frequencies  cause 
partial  or  complete  resonances  in,  and  consequent  reradiation  and  induction  fields  from, 
nearby  metallic  structures.  These  fields  generally  impart  90°  phase  components  (called 
quadrature  effects)  to  the  arriving  wave  fronts,  altering  their  apparent  arrival  direction 
by  amounts  depending  on  the  positions  and  sizes  of  the  resonating  structures.  These 
changes  in  direction  result  in  deviations  of  the  observed  bearing  from  the  true  bearing. 
Deviation,  then,  is  defined  as  the  quantity  that  "must  be  added  algebraically  to  the 
observed  bearing  in  order  to  make  it  equal  the  correct  bearing";  the  deviation,  however, 
does  not  affect  the  resolving  power  (angular  sensitivity)  of  the  DF.  The  quadrature 
effect,  on  the  other  hand,  causes  an  elliptical  polarization  condition  of  the  E  or  H  fields, 
or  both,  which  generally  results  in  failure  of  the  antenna  to  find  a  sharp  null  coupling  posi- 
tion to  these  fields,  thus  adversely  affecting  the  resolution  of  the  DF.  Figure  2  (a)  shows 
the  maximum  deviation  that  may  be  expected  due  to  induced  currents  at  frequencies  at 
which  the  structure  may  be  resonant  (worst  case).  For  frequencies  off  resonance  and 
/<0.5  the  shapes  of  the  curves  of  Fig.  2  (a)  are  maintained,  but  the  magnitude  of  the  devia- 
tion is  reduced  approximately  as  follows: 


D 


D  =  ^*: *  (2) 

1  +  /<?[!  -  (/o//)T 


where  e  is  the  voltage  induced  in  the  structure  at  frequency,  /;  r,  x,  L,  C,  and  Q  are  the 
electrical  constants  of  the  structures;  Z>res.  is  the  maximum  deviation  at  the  resonant 
frequency /0;  and  D  is  the  maximum  deviation  at  any  frequency.  The  resonant  frequency, 
/o,  of  structures  may  be  determined  from  Fig.  2(6),  to  which  some  corrections  should  be 


22-40 


AIDS  TO  NAVIGATION 


._  Hr_  Reradiated  field  .  n   ^Q 

Hd  Direct  field       '     r 

D^stan*1  (F;)=  greatest  possible  deviation 
H=overall  height  of  a  %  wave  (grounded  0,  H)  or  %  wave  (dlpoFe  H,-H)  reradlafor 

1 


F~ 


F=.0125  F=.015 
D=.70°  JD4.85 
10H  '  ' 


Note: 

Solid  curves  use  0-9H  abscissa 
Dotted  curves  use  0-90H  absc  ssa 


-H 


10H 


20H 


60H 


70H 


30H  40H  50H 

Spacing  from  reradiator 
FIG.  2 (a).     Bearing  Deviations  Due  to  Presence  of  a  Reradiator 


80H 


SOH 


Notes.  1.  D  (Worst  Possible  Deviation)  is  directly  dependent  upon  the  lateral  and  vertical  spacings 
from  the  reradiator.  2.  A  resonant  reradiator  of  Q  =  20  has  been  assumed.  3.  For  other  Q's  and 
Bon-resonant  conditions,  see  text.  4.  Curves  show  reradiator  spacings  to  maintain  constant  D  or  F. 


36 
34 
32 
30 
28 
26 
24 
22 

=  16 
14 
12 
10 
8 
6 
4 
2 

4         2          4        A 

X 

T 

T 
A 

4 

\ 

\ 

T 

HX 

4 
HS. 

A 
HX: 

Base 

492 
~    FMO 
_  246 

~         ^MC 

f.et 

\          ^ 

\ 

\ 

\ 

\ 

\ 

feet 

\ 

\ 

\ 

\ 

\ 

\ 

=  -=  —  feet 

jd  oo  velocity 
In  air 

\ 

\ 

s 

\ 

\ 

\ 

\ 

\ 

\ 

\ 

\ 

\ 

\ 

\ 

\ 

\ 

\ 

\ 

\ 

\ 

N 

\ 

\ 

\ 

\ 

\ 

\ 

s 

s^ 

\ 

S 

x, 

"^ 

\ 

^ 

v^ 

^ 

v^ 

<^ 

^ 

"^^ 

-  —  . 

0      5     10    15    20    25    30    35    40    45    50    55    60     65 
"H"  in  feet 

FIG.  2(6).     Fundamental  and  Harmonic  Resonant  Frequencies  of  Conductors  of  Physical  Heights  H 


ESTABLISHED  NAVIGATIONAL  AIDS 


22-41 


added  for  end  effects  and  top  loading  as  shown  in  Fig.  2(c).  The  range  of  Q's  is  approxi- 
mately as  follows: 

Wide-width  reradiators  (superstructures) 1.5-2 

Medium-width  reradiators  (funnels) 2    -3 

Narrow-width  reradiators  (masts) 5     -10 

Very  thin  reradiators  (guy  wires,  antennas) 10     -20 

It  can  be  seen  from  these  results  that  DF  antenna  locations  on  top  of  the  tallest  structures 
give  the  least  reradiation  error  effect  from  those  structures  and,  generally,  maximum 
clearance  (least  error)  from  the  other,  surrounding  structures.  Breaking  up  structures 


-4V- 


}< w ^ 

Ex.  2   Bridge:  < 


T 


tz        -*••* ^— ft         -^ Jr-fy 

Ex.  3(a)  Mast:  d<<W       Ex.  3(6)     Mast  with  yardarm     Ex.  3(c)     Mast  with  loop 


Ex.  4(a)  Tall  mast:  eZ«W              Ex.  4(6)  Tall  mast                  Ex.  4(c)   Tall  mast 
H  >W  with  yardarm with  loop    v 


Note:  r  shows  region  of  strong  reradlated  fields 
FIG.  2(c).     Top  Loading  and  End  Effect  Corrections  for  Shipboard  Reradiator  Heights  17 

(e.g.,  guy  wires,  rails)  with  insulators,  to  reduce  the  induced  currents,  further  reduces  the 
environmental  effects.  Partial  compensation  of  environmental  effects  can  often  be  ob- 
tained by  judiciously  introducing  compensator  loops  or  structures,  such  that  an  approxi- 
mately equal  and  opposite  (compensating)  environmental  effect  is  obtained.  Successful 
means  have  been  devised  for  altering  the  frequency  characteristics  of  structures  surround- 
ing a  DF  antenna  in  order  to  reduce  and  control  environmental  effects.  (See  reference  5.) 
Quadrature  effects  are  generally  reduced  by  minimizing  the  environmental  effects  as 
stated  above.  However,  in  practical  direction  finders,  some  polarization  effects  and  resid- 
ual environmental  effects  often  leave  high  values  of  quadrature  effect  which  must  be  further 
minimized  in.  order  to  realize  satisfactory  DF  bearing  resolution.  Quadrature  balance  or 
compensation  has  long  been  used  for  this  purpose.  It  makes  use  of  a  non-directional 
antenna  for  obtaining  a  voltage  from  the  arriving  wave  which,  with  a  properly  adjusted 
phase  and  amplitude,  can  be  made  to  cancel  out  completely  the  undesired  quadrature 
voltage  derived  from  the  directional  collector.  In  practice,  the  phase  shift,  if  any,  required, 
for  the  above  balancing  or  compensating  voltage  is  a  constant  value  over  the  frequency 
band.  As  a  result,  most  direction  finders  operating  below  2000  kc/s  incorporate  this 
balance  circuit  with  only  a  single  associated  panel  control  for  positive  and  negative  magni- 
tude of  compensation  adjustment.  The  non-directional  antenna  employed  for  balancer 
purposes  is  actually  the  same  one  generally  furnished  in  practically  all  direction  finders  for 


22-42 


AIDS  TO  NAVIGATION 


±24- 
M 


FIG.  3. 


(a) 


Vector  Relations  for  Quadrature-effect  Suppression  by  Mod- 
ulation Method 


sense  determination,  i.e.,  for  resolving  the  180°  ambiguity  in  the  direction  of  a  received 
signal.  When  used  for  that  purpose,  the  phase  and  magnitude  of  the  sense  antenna  voltage 
is  adjusted  (generally  by  a  fixed  circuit)  to  match  (as  closely  as  possible)  the  phase  and 
magnitude  of  the  directional  antenna's  maximum  coupling  output.  In  this  manner  the 
cosine  (double  null)  coupling  law  is  converted  to  an  approximate  cardioid  (single  null) 
coupling  law  which  possesses  the  desired  single  non-ambiguous  null.  Because  the  cardioid 
law  yields  a  poorer  bearing  resolution,  i.e.,  differential  amplitude  to  angle  ratio,  in  the 
region  of  its  null  as  compared  to  the  cosine  law  null  region,  the  cardioid  is  rarely  used  for 
other  than  sense  determination. 

Receiver-introduced  modulation  for  quadrature-effect  suppression  (see  reference  6), 
which  has  recently  been  developed,  possesses  several  advantages  over  the  balancer  type. 
It  operates  on  the  principle  of  phase  discrimination  for  the  undesired  quadrature  com- 
ponent. Phase  discrimination  is  accomplished  by  introducing  the  voltage  e\,  derived 
from  the  non-directional  antenna,  into  the  DF  as  a  reference  voltage  of  proper  amplitude 
and  phase  with  respect  to  the  directional  voltage,  e-2.  The  directional  voltage  ez  is  then 
passed  through  a  mechanical  or  electronic  reversing  switch  just  before  it  is  added  to  e\. 
The  resultant  e*  has  a  maximum  +es  and  a  minimum  —  63  value  depending  upon  the 
position  of  the  reversing  switch  as  shown  in  Fig.  3 (a).  It  will  be  noted  in  Fig,  3(6)  that 

the  undesired  or  quadrature 
directional  antenna  voltage 
component  e*',  after  pass- 
ing through  the  common 
reversing  switch,  is  also  vec- 
torially  added  to  e\.  The 
resultant  e$  thus  also  has 
values  +63'  and  —63'  de- 
pending upon  the  position  of 
the  reversing  switch.  How- 
ever, it  will  be  noted  that 
the  desired  directional  volt- 
age ±62  causes  a  maximum 
differential  effect  on  63,  whereas  the  undesired  or  quadrature  directional  voltage  ±62' 
causes  a  minimum  (zero)  differential  effect  on  e$f.  Moreover,  for  small  values  of  quad- 
rature, the  magnitude  of  +es'  differs  negligibly  from  e\>  Thus,  if  e^  is  10  per  cent  of  e\, 
±e3  =  V(l)2  +  (O.I)2  =  1.005,  or  the  presence  of  10  per  cent  62'  bauses  only  0.5  per  cent 
increase  in  63  over  ei.  By  utilizing  suitable  synchronous  commutators, .  or  sweep  circuits 
with  a  cathode-ray  tube,  the  maximum  and  minimum  resultant  voltages  ±63  can  be  com- 
pared and  the  desired  sharp  null  position  of  the  collector  can  be  obtained  just  as  readily 
with  considerable  quadrature  voltage  present  as  when  none  is  present,  since  the  entire 
quadrature  suppression  is  continuous,  automatic,  and  independent  of  the  operator.  It 
should  be  noted  that  positive  sense  determination  is  continuously  present  in  this  type 
of  system  since  the  non-directional  reference  voltage,  ei,  is  automatically  a  sense  voltage, 
and  sense  is  obtained  merely  by  correlating  the  instantaneous  position  of  the  reversing 
switch  with  an  increasing  (or  decreasing)  resultant  output.  It  can  be  shown  that  the 
receiver-introduced-modulation  principle  markedly  improves  the  signal-to-noise  ratio 
(or  the  bearing  sensitivity)  by  integrating  out  the  random  (non-synchronous)  noise  effects. 
It  can  be  effectively  demonstrated  that  satisfactory  DF  bearings  can  be  obtained  on  broad- 
cast program  modulated  signals  notwithstanding  noise  levels  that  obliterate  the  program 
intelligence.  The  cathode-ray  tube  comparator  for  "on  bearing"  indication  in  some  U.  S. 
Navy  direction  finders  is  especially  desirable  for  obtaining  indication  on  ICW  transmis- 
sions. When  CW  or  MOW  transmissions  are  employed  the  entire  DF  bearing  may  be 
taken  automatically  by  employing  a  suitable  controlled  servo  system  to  orient  the  direc- 
tional collector  for  minimum  difference  in  H-ea  and  —63.  This  identical  principle  is  em- 
ployed in  the  automatic  direction  finder  (ADF)  described  in  article  3  under  Aids  to  Air 
Navigation,  and  as  a  second  mode  of  operation  in  the  Navy  DBD  MF/DF  system,  the 
latter  being  especially  designed  for  simplified  marine  use.  Both  the  balancing  and  the 
modulator  suppression  of  quadrature  effects  may,  under  certain  conditions,  introduce  a 
deviation  effect.  This  may  occur  when  a  large  quadrature  (undesirable)  voltage,  e\  ,  is 
induced  in  the  non-directional  antenna.  This  condition,  and  its  amount,  can  be  quickly 
identified  by  a  shift  of  the  reciprocal  bearing  (i.e.,  observed  bearing  plus  180°)  from  its 
normal  180°  position.  Since  this  effect  is  almost  identical  to,  and  originates  from,  the  same 
mechanisms  responsible  for  the  normal  Deviation,  it  is  as  stable  as  the  latter,  and  can  be 
absorbed  by  the  overall  DF  calibration  when  made. 

Calibration  Curves.  A  conventional  DF  calibration  curve  is  shown  in  Fig.  4(a)  which 
is  generally  taken  for  a  few  of  the  most  received  frequencies.  The  same  data  can  be  pre- 


ESTABLISHED  NAVIGATIONAL  AIDS 


22-43 


sented  in  the  automatic  interpolation  (see  reference  5)  form  shown  in  Fig.  4(6) .    The  latter 
form  permits  more  direct  plotting  and  utilization  of  the  calibration  data  with  the  great 


2?  4-1 

r 

> 

(. 

00  1 

N 

<c  ,. 

X 

"\ 

X 

-^ 

"°       0 

A 

V. 
/ 

1  / 

/"   \ 

\ 

^ 

^ 

00  k 

c 

/  / 

/^ 

\ 

^ 

c      u 

\ 

ss. 

s 

/ 

\ 

s 

/ 

V,. 

o-l 

\ 

\  " 

/ 

'•5-2 

\ 

1 

\ 

\ 

1 

/ 

Q 
-3 

V 

y 

^ 

J 

C 

) 

4 

5 

9 

0 

i: 

35 

It 

30 

2: 

25 

2; 

rQ 

31 

5 

36 

Observed  bearing 
FIG.  4  (a).     Conventional  Deviation  Chart  for  Direction-finding  Calibration 

advantage  that  the  corrected  bearing  of  other  than  calibrated  frequencies  may  be  ascer- 
tained just  as  conveniently  as  that  of  the  calibrated  frequencies. 

Polarization  Effects.     Radio  waves  propagated  with  their  electric  field  vertical  have 
become  known  as  vertically  polarized  waves;  those  having  their  electric  field  horizontal 

D-F  calibration  Interpolation  charts 
fore-starboard  quadrant 


iooh 


3551- 


1.5 


2.0  2.5  3.0 

Frequency-megacycles 
FIG.  4(6).     Automatic  Interpolation  Chart  for  Direction-finding  Calibration 


4,0 


have  become  known  as  horizontally  polarized  waves;  those  having  their  electric  field  in- 
clined are  a  combination  of  both  types.  Experience  shows  that  most  direction  finders  are 
susceptible  to  appreciable  unwanted  polarized  energy  pickup,  notwithstanding  the  care 


22-44 


AIDS  TO  NAVIGATION 


taken  to  exclude  it  from  the  antenna  and  feeder  system.    For  a  discussion  of  this  effect 
and  its  cure  see  Section  6,  article  32. 

WEATHER  AND  TIME  TRANSMISSIONS.  Radio  navigational  warnings  and  stand- 
ard time  signals  are  available  to  all  ships  equipped  with  communication  receivers.  The 
navigational  warnings  include  the  local  weather  forecast  plus  any  urgent  information  with 
regard  to  tidal  waves,  offshore  winds,  ice,  and  storms. 

10.  RECENTLY  INTRODUCED  NAVIGATIONAL  AIDS 

RADAR.  (See  references  7-12.)  Present  radar  systems  provide  one  of  the  few  known, 
yet  most  reliable,  methods  for  surface-obstacle  detection  under  conditions  of  restricted 
visibility,  whether  these  obstacles  are  other  ships,  icebergs,  buoys,  islands,  landmarks,  etc. 
Radar  provides,  in  effect,  an  electronic  searchlight  aboard  ship,  capable  of  "seeing" 
through  darkness  or  fog,  in  any  weather  condition,  for  ranges  up  to  approximately  25-30 
miles  (line  of  sight),  depending  on  the  power  of  the  "searchlight."  The  use  of  radar, 
therefore,  will  be  most  applicable  to  collision  prevention  at  sea,  iceberg  detection,  harbor 
navigation,  coastal  navigation,  and  harbor  control  from  shore  stations. 

Table  2  lists  minimum  specifications  for  marine  navigational  radar.  These  specifications 
suggest  the  use  of  PPI  (plan  position  indication)  presentation  of  the  echo  information 
(see  Section  15,  article  24). 

Table  2.    Performance  Factors  for  Some  Marine  Navigational  Radars 


Performance 
Factors 

TJ.S.C.G. 
Class  A  Spec.* 

Manufac- 
turer A 

Manufac- 
turer B 

U.S.C.G. 
Class  B  Spec.f 

Manufac- 
turer C 

Manufac- 
turer D 

Maximum  range.  .  . 
Minimum  range.  .. 
Range  resolution 

30  miles 
100yd 
100yd 

32  miles 
100yd 
100yd 

40  miles 
100yd 
100yd 

30  miles 
400yd 
200  yd 

30  miles 
200yd 
100yd 

50  miles 
100yd 

Bearing  resolution 

4°  (on  10  cm) 

3° 

3° 

6° 

3° 

Presentation 

3°  (on  3  cm) 
1"  PPI 

7"  PPI 

12"  PPI 

1"  PPI 

1"  PPI 

7"  PPI 

Range  scales 
(miles) 
Range  acfiUTAcy 

2-5,  4-15,  and 
15-30 
±2%  or  ±50 

2,  8,  and  32 

2,  10,  and  40 

±2% 

2-5,  4-15,  and 
15-30 
±2%  or  ±100 

2,  6,  and  30 
±2%  or  ±100 

1.5,5,  15,  and 
50 

±2% 

Bearing  indicator.  . 

Antenna: 
Beam  width  

Rotation  
Frequency  

yd 
True 

4°H,  15°V(on 
10  cm);  2°  H, 
15°  V  (on  3 
cm) 
360°,  6-15  rpm 
3000-3246 

Relative  true 
(avail.) 

2°  H,  15°  V 
360°,  12  rpm 

True 
2°H,  15°  V 

360°,  15  rpm 

yd 
True  or  relative 

5°H,  15°  V 

360°,  6-  15  rpm 
3000-3246 

yd 
True 

5°H,  15°  V 

360°,  1  1  rpm 
3200 

Relative  true 
(avail.) 

3.5°  H,  12°V 

360°,  7  rpm 
3070  ±  50 

(Me/s) 

9320-9500 

9320-9430 

9320-9430 

9320-9500 

R-f  source  

Magnetron 

Magnetron 

Magnetron 

Magnetron 

Magnetron 

Peak  power  

15  kw 

15  kw 

35  kw 

7   kw    (on    10 

7kw 

15  kw 

Pulse  rate  
Pulse  width  

Receiver: 
Band  width 

800  cps 
0.5   MS    (maxi- 
mum) 

2000  cps 
0.4  MS 

800  cps 
0.25  MS 

8Mcs 

cm),    15   kw 
(on  3  cm) 
800  pps 
1  .0   MS    (maxi- 
mum) 

Optimum 

1500  pps 
0.5^8 

Optimum 

1000  pps 
0.4  MS 

3  Mcs 

Gain  

120  db 

120db 

117  db 

*  Class  A  corresponds  to  U.S.C.G.  Minimum  Specifications,  Brief  No.  i;  Nov.  14,  1945  (revised  Aug.  1,  1946). 
t  Class  B  corresponds  to  U.S.C.G.  Minimum  Specifications,  Brief  No.  2,  Nov.  14,  1945  (revised  Aug.  1, 1946). 

Principles.  A  short  powerful  burst  of  electromagnetic  energy  is  emitted  at  a  known 
spot  and  is  narrowly  beamed  in  a  given  known  direction.  Returning  echoes  from  ob- 
jects within  an  arbitrary  range  in  that  given  direction  are  received  at  the  known  spot, 
detected,  and  visually  displayed.  For  persistence  and  continuity  of  display,  the  emitted 
pulses  are  repeated  periodically  at  a  fixed  rate  with  enough  intervening  time  to  allow  the 
return  of  any  echoes.  If  an  echo  from  an  object  is  received  by  the  system  after  a  time 
delay,  T,  from  the  initial  burst,  the  distance  of  the  object  from  the  radar  system  is: 


(3) 


RECENTLY  INTRODUCED  NAVIGATIONAL  AIDS       22-45 

where  c  is  the  velocity  of  light  (or  electromagnetic  waves)  in  air  and  where  the  time  T  is 
measured  from  the  beginning  of  the  transmitted  burst  to  the  beginning  of  the  received 
echo.  Thus,  the  distance  to  the  object  is  determined.  If  the  antenna  system  is  sufficiently 
directive,  the  pointed  direction  of  the  antenna  at  the  time  an  echo  is  received  is  the  direc- 
tion of  the  echo,  thus  furnishing  a  bearing  determination. 

^  Unless  the  wavelength  of  the  radiations  used  is  small  compared  with  the  linear  dimen- 
sions of  the  reflecting  object,  the  phenomenon  of  diffraction  takes  place,  making  the  echo 
amplitude  inversely  proportional  to  the  square  of  the  wavelength.  If  the  wavelength  is 
small  compared  with  the  reflecting  object,  the  amplitude  of  the  echo  field  does  not  sensibly 
depend  upon  the  magnitude  of  the  wavelength  but  rather  upon  the  nature  of  the  reflecting 
object.  In  the  case  of  the  free-space  propagation  between  the  radar  system  and  the 
reflecting  object,  the  following  relationship  holds: 


where  Pr  is  the  received  power  at  the  receiver  input  terminals  (watts),  AQ  is  the  effective 
absorption  cross-section  of  the  receiving  antenna  (square  meters)  ,  Go  is  the  overall  power 
gain  of  the  feeder  and  the  radiating  antenna,  Pt  is  the  transmitter  peak  power  (watts),  K 
is  a  complex  reflection  coefficient  dependent  on  the  nature  of  the  target  and  is  given  as 
the  effective  echo  area  of  the  target  in  the  direction  of  the  radar  (square  meters)  ,  r  is  the 
distance  in  meters  from  the  radar  to  the  target.  The  second  form  of  the  equation  is  in 
terms  of  the  maximum  range,  rmax.,  and  the  minimum  detectable  power  received,  Pmin.- 
These  results  for  free  space  propagation  must  be  modified  for  propagation  over  a  spherical 
earth  since,  in  the  range  of  frequencies  used,  electromagnetic  waves  are  propagated  over 
approximately  a  "line  of  sight"  path  with  small  diffraction  effects  occurring  at  the  lower 
radar  frequencies.  Thus,  the  antenna  height  above  the  sea,  the  target  height  above  the 
sea,  and  the  height  of  any  intervening  objects  must  be  taken  into  account. 

From  these  considerations,  it  is  evident  that,  in  order  to  utilize  efficiently  the  principles 
of  reflection  with  electromagnetic  waves,  a  radar  system  must  (1)  generate  a  wave  whose 
length  is  small  compared  with  the  objects  from  which  the  wave  is  to  be  reflected;  (2) 
generate  enough  power  at  that  frequency  to  be  able  to  receive  and  detect  the  returning 
signal;  (3)  provide  a  means  of  measuring  the  tune  delay  from  the  transmission  to  the  recep- 
tion of  that  signal. 

Fundamental  System  Constants.  Each  radar  system  has  associated  with  it  certain 
constants  whose  choice  depends  upon  the  available  components,  the  desired  operational 
performance,  and  the  intended  use  of  the  system.  The  normal  variations  of  these  constants 
are  as  indicated  in  the  following: 

(a)  Carrier  Frequency.  The  choice  of  carrier  frequency  depends  on  the  permissible 
dimensions  of  the  antenna  system  to  -be  used  and  the  directivity  or  beam  sharpness  desired, 
since  the  size  of  the  antenna  system  reduces  with  increased  frequency,  and  the  directivity, 
as  well  as  gain,  improves  with  frequency  for  a  given  antenna  size.  Thus  the  lower  fre- 
quency is  limited  by  the  practical  antenna  size;  the  upper  frequency  is  limited  by  atmos- 
pheric reflection  and  absorption  effects  (pronounced  near  3000  Me  or  10  cm  wavelength 
for  reflection  and  30,000  Mc/s  or  1  cm  wavelength  for  absorption)  and  the  availability  of 
tubes  capable  of  generating  and  amplifying  enough  radio-frequency  energy  to  provide 
the  necessary  range.  The  lower  frequency  limit  is  about  100  Me,  though  frequencies  higher 
than  30,000  Me  have  been  successfully  used  experimentally. 

(6)  Transmitter  Pulse  Width,  The  minimum  range  at  which  an  object  can  be  detected 
by  a  radar  system  is  determined  largely  by  the  width  of  the  transmitted  pulse  (at  the 
x/2  power  point)  since  an  echo  returning  while  the  transmitter  is  still  operating  will  be 
masked  by  the  transmitter  pulse.  This  is  even  more  true  if  the  receiver  is  always  blocked 
or  desensitized  for  the  duration  of  the  transmitter  pulse.  The  usual  pulse  widths  range 
from  0.25  to  2.0  /-is  for  navigational  purposes. 

(c)  Pulse  Repetition  Rate.    The  pulse  repetition  rate  must  be  slow  enough  to  allow  time 
for  the  maximum  range  echoes  to  return  to  the  antenna  before  another  pulse  is  transmittedf 
and  it  must  be  fast  enough  to  provide  enough  traces  while  the  antenna  is  rotating  or 
pointing  in  a  given  direction  to  produce  a  lasting  indication  on  an  oscilloscope  screen. 
Therefore  the  maximum  range  determines  the  highest  pulse  rate,  and  the  rotational  speed 
of  the  antenna  determines  the  lowest  pulse  rate  that  can  be  used.    In  practice  these  rate* 
vary  from  less  than  60  to  several  thousand  pulses  per  second. 

(d)  Duty  Cycle.    The  duty  cycle  of  a  radar  system  is  the  ratio  of  the  average  power  to 
the  peak  pulse  power.     It  depends  on  the  relation  between  pulse  width  and  the  pulse- 
repetition  time.    Thus  a  lower  duty  cycle  permits  higher  peak  power  operation  at  the  same 
average  power.    The  maximum  range  of  the  system  is  dependent  on  the  peak  power  for  a 
given  pulse  width,  while  low  average  power  means  smaller  tubes  and  components  ia  the 


22-46 


AIDS  TO   NAVIGATION 


,-Awir- 


transmitter.  The  limit  to  peak  power,  however,  for  a  given  transmitter  tube  is  the  break- 
down potentials  between  the  various  electrodes.  In  practice,  radar  duty  cycles  vary 
between  0.005  and  0.0001. 

Fundamental  System  Components  (General).     Radar  systems  now  in  existence  differ 
-widely  in  detailed  design  and  complexity,  depending  on  their  functional  use  and  the  accu- 
racy and  amount  of  information  required.    However,  a  single  basic  system  can  be  visualized 
in  which  the  functional  requirements  fit  equally  well  almost  all  specific  requirements. 
The  six  primary  components  are  shown  in  Fig.  5;  they  may  be  summarized  as  follows: 
(a}  Timer  or  Synchroniser.    The  timer  determines  the  pulse-repetition  rate  of  the  radar 
and  provides  a  zero  reference  point  for  time  measurements  and  for  operation  of  sequential 
functions  in  a  definite  time  relationship.    Such  timing  may  be  supplied  by  a  separate  unit 

such  as  a  sine-wave  oscillator, 

T  y  a  multivibrator,  or  a  blocking 

oscillator  with  the  necessary 
pulse-shaping  circuits.  An- 
other commonly  used  method 
of  timing  is  to  make  the 
transmitter  with  its  associ- 
ated circuits  establish  its  own 
repetition  rate  and  provide 
the  synchronizing  pulses  for 
the  rest  of  the  system.  This 
action  may  be  accomplished 
by  a  self-pulsing  or  blocking 
radio-frequency  oscillator,  or 
by  a  rotating  spark  gap. 
Self-timing  eliminates  a  num- 
ber of  special  timing  circuits, 
but  the  pulse  width  and  pulse- 


FIQ.  5.     Functional  Block  Diagram  for  Basic  Radar  System 


repetition  rate  are  not  usually  as  stable  or  rigidly  controllable  as  is  necessary  for  some 
applications. 

(b)  Transmitters.    To  generate  pulses  of  high-frequency  electromagnetic  waves  at  high 
power  levels  a  conventional  pulse  transmitter  is  used.    Care  must  be  taken  that  the  tubes 
are  suitable  not  only  for  the  average  power  dissipation  but  also  for  the  high  powers  and 
voltages  during  the  pulse. 

(c)  Antenna  System.    The  purposes  of  the  antenna  system  are  to  beam  and  radiate  the 
energy  efficiently  from  the  transmitter  into  space,  to  focus  and  pick  up  the  returning  echo 
and  pass  it  on  to  the  receiver.    A  transmit-receive  switch  (TR  box)  must  be  included  to 
prevent  the  transmitter  energy  from  harming  the  receiver. 

(d)  Receiver.    A  conventional  wide-band  receiver  is  used. 

(e)  Indicator.    To  display  the  detected  pulses  visually  so  that  range,  bearing,  or  eleva- 
tion of  any  echo  source,  or  combination  of  these,  can  be  determined.     See  Section  15, 
article  24. 

(/)  Power  Supply.  A  conventional  well-regulated  low-impedance  power  supply  is  used. 
Pulse-forming  lines  build  up  a  charge  during  the  inactive  period  to  store  energy  to  discharge 
in  the  pulse. 

Performance  Factors,  (a)  Resolution.  The  resolution  of  obstacles  by  a  radar  system 
will  depend  on  the  pulse  width,  the  effective  antenna  beam  width,  the  receiver  band  width, 
the  frequency,  and  the  stability  of  the  entire  system.  For  a  typical  commercial  naviga- 
tional radar  in  the  3-cm  (10,000-Mc)  band,  having  a  receiver  band  width  of  8  Mc/s, 
antenna  beam  width  of  2°  (to  half  power  points),  and  a  pulse  width  of  1/4  //s,  the  resolution 
is  100  yards  in  range  and  3°  in  azimuth.  Above  approximately  1000  Mc/s  the  frequency 
is  not  a  major  determining  factor  in  itself,  but  only  as  it  affects  the  other  factors,  since  the 
•wavelength  is  small  enough  to  provide  efficient  reflection  from  the  smallest  objects  ordi- 
narily encountered. 

(b)  Maximum  and  Minimum  Range.  The  maximum  range  will  depend  on  the  height  of 
the  antenna  system  above  the  sea,  the  power  output  of  the  transmitter,  and  the  gain  and 
efficiency  of  the  antenna  system.  An  average  ship  installation  would  have  a  maximum 
range  of  from  25—30  miles,  which  is,  in  general,  great  enough  for  the  intended  navigational 
use.  -  The  minimuin  range  will  depend  directly  on  the  pulse  width  and  recovery  time  of 
the  receiving  system  (including  antenna  switching  mechanism) ;  for  the  typical  commercial 
system  mentioned  above,  it  is  100  yards.  The  radar  indicator  is  usually  provided  with  a 
range  switch  permitting  any  of  several  discrete  ranges  to  be  displayed  on  the  screen.  The 
accuracy  of  the  range  information  depends  on  the  initial  accuracy  and  stability  of  the  range 
markers;  azimuthal  accuracy  usually  depends  on  the  accuracy  of  tracking  between  the 


RECENTLY  INTRODUCED  NAVIGATIONAL  AIDS       22-47 


antenna  rotation  and  the  sweep  rotation  on  the  cathode-ray  tube  face.    Accuracies  of 
2  per  cent  in  range  and  2°  in  azimuth  are  not  unusual. 

(c)  Presentation.  The  most  usable  form  of  the  radar  information  for  marine  navigational 
purposes  is  the  PPI  type  presentation  on  a  screen  of  5  to  9  in.  in  diameter,  with  provision 
for  true  or  relative  bearing  stabilization,  range  markers,  and  range  scale  selector.  The  ratio 
between  different  range  scales  is  often  made  the  same. 

(d")  Installation,  Maintenance,  and  Operation.  The  present  radar  systems  are  designed 
for  operation  by  bridge  personnel  having  little  or  no  technical  training.  The  indicator  is 
mounted  in  the  pilot  house,  and  the  antenna  is  properly  located  to  provide  360°  clearance 
to  the  horizon.  To  facilitate  this  arrangement  on  all  types  of  vessels,  the  radio-frequency 
components,  the  antenna  assembly,  and  the  indicator  are  usually  manufactured  as  separate 
units. 

RADAR-BEACONS  (RACON).  (See  references  10-12.)  Radar  piloting  is  beset  by  two 
important  problems.  First,  targets  for  navigational  purposes  are  sometimes  small  or 
poorly  denned,  resulting  in  small  or  weak  echoes,  especially  near  the  maximum  range  of 
the  radar  system.  Secondly,  these  weak  echoes  in  the  presence  of  sea  return  often 
leave  the  absolute  identification  of  the  navigational  targets  in  doubt.  These  prob- 
lems are  greatly  simplified,  however,  by  providing  a  system  of  electronic  beacons  designed 
to  serve  as  aids  to  navigation  for  ships  equipped  with  the  proper  radar  equipment.  The 
term  Racon,  a  contraction  of  radar-beacon,  designates  such  a  system.  These  beacons  or 
racons  are  designed  to  emit  or  reflect  a  large  amount  of  energy  which  will  allow  dependable 
target  indication  at  a  much  greater  distance  than  that  obtainable  when  normal  reflections 
alone  are  depended  upon.  There  are  three  types  of  such  beacons,  and  each  is  described 
below: 

Radar  Respondor-beacons  (Transponder).  These  racons  consist  of  transponders,  or 
pulse-type  receiver-transmitters,  located  at  strategic  ground  sites  such  as  coastal  points 
and  islands,  which  receive 
interrogating  signals  from 
a  radar  transmitter  and 
automatically  send  back 
identifying  reply  pulses  to 
the  radar.  The  coded  reply 
appears  on  the  radar  oscil- 
loscope in  such  a  way  that 
both  position  and  identifi- 
cation of  the  racon  station 
are  indicated.  Reference  to 
prepared  code  lists  or  charts 
then  serves  to  associate  that 
respondor-beacon  with  a 
fixed  navigational  point. 
Thus  a  navigator  will  be 
able  to  check  his  course,  or 
even  completely  guide  his 
craft,  since  range,  bearing, 
and  identification  will  be  ob- 
tained from  each  respondor- 
beacon.  A  typical  radar 
and  fixed  station  microwave 


Receiving 
antenna 


Transmitting 
antenna 


Functional  Block  Diagram  of  Basic  Racon  System 


racon  transpondor  operation  is  shown  in  the  block  diagram  in  Fig.  6. 

To  prevent  racon  signals  from  appearing  on  the  radar  screen  all  the  time,  and  to  avoid 
continual  interrogation  of  the  beacon  by  radar,  the  system  is  designed  to  operate  under 
the  following  conditions:  (1)  the  beacon  responds  only  to  pulses  of  2-jus  or  greater  duration 
within  an  appropriate  frequency  band;  (2)  the  beacon  replies  at  a  fixed  frequency  which  is 
common  to  the  entire  beacon  system  and  which  is  just  outside  the  radar  system  frequency 
band.  Ten-centimeter  band  racons  respond  to  2-/zs  challenging  pulses  in  the  3000-3246 
Mc/s  band  and  reply  at  3256  Mc/s;  3-cm  band  beacons  receive  pulses  in  the  9320-9500 
Mc/s  band  and  reply  at  9310  Mc/s.  An  additional  band  near  5  cm  is  being  made  available 
for  radar  and  racon  use.  For  racon  system  use,  the  radar  equipment  is  designed  to  provide 
the  necessary  challenging  pulse  length  and  the  correct  receiver  tuning  upon  activation 
of  a  special  button  or  switch.  The  normal  radar  plot  will  then  disappear  and  only  radar- 
beacon  responses  will  appear  on  the  screen. 

Three  types  of  coding,  to  insure  positive  identification  of  a  given  respondor-beacon, 
have  been  devised:  (1)  Sequence  coding  is  provided  by  emitting  a  number  of  half-second 
transmissions  whenever  the  beacon  is  triggered.  During  the  half  second,  pulses  of  a  given 


22-48  AIDS  TO  NAVIGATION 

duration  appear  as  a  pip  of  a  certain  width  on  the  scope.  However,  the  pulses  making  up 
the  next  transmission  may  all  be  of  longer  duration,  thus  giving  a  pip  of  greater  width  on 
the  scope.  The  complete  code  is  made  up  of  the  number  and  combination  of  widths  of 
these  pips.  (2)  Gap  coding  provides  periodic  interruption  of  a  series  of  pulses.  During 
each  period  of  interruption,  identification  is  given  by  a  pulsed  dot  and  dash  transmission. 
Both  sequence  and  gap  coding  are  slow,  requiring  considerable  time  for  complete  identi- 
fication. (3)  Range  coding  gives  an  immediate  identification  as  the  complete  code  is 
produced  on  the  oscilloscope  at  once.  Each  reply,  caused  by  one  interrogation  pulse, 
consists  of  six  pulses  which  appear  on  the  PPI  at  different  positions  along  a  given  bearing 
with  the  first  pulse  indicating  the  beacon  position.  Such  coding  is  accomplished  by  chang- 
ing the  spacing  of  the  pulses. 

Separate  broadside  arrays  serve  as  receiving  and  transmitting  antennas  for  the  trans- 
ponder. Both  are  non-directional  in  the  horizontal  plane  in  order  to  respond  to  interro- 
gations from  any  point  on  the  compass,  but  have  a  narrow  low-angle  vertical  pattern  to. 
facilitate  long-range  operation.  From  the  receiving  antenna  the  received  radar  signals 
go  to  either  a  crystal-video  or  superheterodyne  receiver  which  must  satisfy  the  follow- 
ing requirements:  (1)  the  radio-  and  intermediate-frequency  stages  must  be  broad-band 
in  order  to  receive  signals  at  any  frequency  within  a  given  radar  band  (10-cm  or  3-cm,  but 
not  both);  (2)  the  video  stages  must  not  excessively  widen  1-jus  radar  pulses;  (3)  the  re- 
ceiver must  not  be  easily  blocked;  (4)  the  amplification  must  be  adequate  for  the  received 
radar  signals.  The  output  of  the  receiver  goes  to  a  recoder  which  consists  of  a  discrim- 
inator, coder,  and  gate  multivibrator.  The  discriminator  accepts  variable  amplitude- 
pulses  of  2-/IS  or  greater  duration  but  rejects  all  pulses  of  shorter  duration.  The  coder 
(in  this  case,  a  range  coder)  consists  of  six  start-stop  multivibrators  whose  outputs  are- 
combined  in  an  amplifier  to  produce  a  series  of  pulses.  The  code  may  include  a  maximum 
of  six  pulses  with  15-  or  35-Ats  spacings  between  successive  pulses.  The  gate  consists  of  a 
175-jus  negative  rectangular  pulse  that  is  fed  back  to  the  discriminator,  where  it  prevents 
the  coder  from  accepting  other  receiver  video  signals  until  after  the  completion  of  one 
range-coded  transmission.  The  pulses  from  the  coder  are  applied  to  a  modulator  where 
they  are  converted  into  very  large  amplitude  rectangular  pulses  of  Va-MS  duration.  The 
modulator  triggers  a  transmitting  magnetron  that  oscillates  at  the  proper  reply  frequency. 
To  insure  operation  exactly  on  the  radar  transmission  frequency,  an  automatic  frequency 
control  circuit  is  used  to  keep  the  magnetron  in  tune.  Minimum  peak  powers  of  10  kw 
are  used,  while  some  racons  operate  at  peak  powers  as  high  as  40  kw. 

Transponder  beacons  of  the  type  described,  and  radar  equipments  which  use  them, 
must  be  designed  as  "sister"  equipments,  as  each  is  dependent  upon  the  other.  For  this 
reason  the  Coast  Guard's  specifications  for  commercial  radar  equipment  recommended 
that  provisions  be  made  for  convenient  future  modifications  of  the  radar  for  use  with  racons 
or  microwave  beacons.  Considerable  investigation  and  development  work  has  been  under 
way  to  permit  adaptation  of  the  wartime  racon  system  to  general  marine  navigational 
purposes. 

RADAR  MARKER  BEACONS  (RAM ARK).  A  second  type  of  radar  beacon  consists 
of  a  simple  shore-based,  continuously  pulsed  beacon  operating  on  a  single  frequency. 
Rather  than  being  interrogated  by  the  radar  pulses,  this  type  continuously  transmits, 
and  the  radar  need  have  only  a  means  for  receiving  the  beacon  frequency  when  it  is  desired 
to  utilize  the  beacon.  This  eliminates  the  necessity  for  sending  out  2-jus  pulses  and  simpli- 
fies the  ground  equipment  so  that  it  acts  in  much  the  manner  of  the  present  low-frequency 
radiobeacons.  The  navigator  sees,  on  his  PPI,  a  line  extending  radially  from  the  center  to 
the  periphery  of  the  scope  on  the  bearing  of  the  beacon.  Two  beacons  are  needed  to  obtain 
a  fix  since  no  range  information  is  provided.  The  means  of  identification  is  the  location 
of  the  aid  with  respect  to  the  naturally  distinguishable  land  contours  shown  on  the  radar. 
Additional  means  of  identification  have  been  considered  also,  e.g.,  groups  of  pulses,  known 
repetition  rates,  or  modulation  of  the  signal  with  an  audio  code,  but  these  require  addi- 
tional modifications  to  the  radar. 

RADAR  REFLECTOR  BEACONS  (CORNER  REFLECTORS).  A  third  type  of  beacon 
classed  as  a  racon  is  the  "corner  reflector,"  which  is  merely  a  mechanical  assembly  of 
sheet-metal  faces,  designed  to  give  the  maximum  reflection  of  energy  received  from  any 
given  direction  over  fairly  wide  angular  limits.  The  reflector  is  inexpensive  and  easy  to- 
construct  and  maintain.  These  devices  have  no  means  of  identification  other  than  their 
prominent  echoes  on  the  radar  screen  and  the  fact  that  their  positions  can  be  plotted  on. 
charts.  The  great  simplicity  and  economy  of  such  a  scheme  suggest  its  use  wherever 
feasible.  In  rough  weather,  however,  it  may  be  very  difficult  to  distinguish  its  return 
through  the  ground  clutter. 

LOR  AN.  (See  references  11,  13,  and  14.)  The  accelerated  development  of  radar  and 
other  electronic  equipments  during  World  War  II  has  reduced  to  relatively  simple  opera- 


RECENTLY  INTRODUCED  NAVIGATIONAL  AIDS       22-49 


Loran  line  of  position, 

(hyperbola  of  constant 

time  difference^ 800  jus) 


tional  procedures  the  accurate  measurement  of  radio  transmission  propagation  time.  This 
has  resulted  in  the  introduction  and  wide  use  of  two  important  radio  position-determining 
systems  utilizing  this  propagation  time  principle,  i.e.,  Loran  and  Gee.  In  these  systems,  a 
position  fix  is  obtained  from  the  intersection  of  two  loci  of  position,  each  locus  of  position 
being  determined  by  measuring  the  propagation  time  difference  between  two  synchronized 
pulsed  signal  emissions  arriving  from  two  known  but  widely  spaced  radio  transmitting 
sources.  Since  the  velocity  of  propagation  of  radio  waves  over  the  earth's  surface  is 
essentially  constant,  propagation-time-difference  measurement  varies  directly  as  the 
distance  difference  between  a  receiving  station  and  two  fixed  transmitters.  Therefore, 
the  locus  of  points  which  are  a  given  constant  time  difference  from  two  fixed  points  is  a 
hyperbola  with  the  fixed  points  as  foci.  Thus  the  two  transmitting  stations  or  fixed  points 
provide  a  family  of  hyperbolas  about  the  stations,  each  hyperbola  representing  a  constant 
value  of  propagation  time  difference.  Position-fixing  methods  which  use  such  a  family  of 
hyperbolas  can  be  classed  as  hyperbolic  systems,  and,  in  addition  to  Loran  and  Gee,  they 
include  the  Dingley  system,  Decca,  and  Popi.  These  last  two  systems  utilize  hyperbolas 
derived  from  constant  propagation  phase  differences  (identical  in  principle  to  optical 
interference  phenomena)  instead  of  constant  propagation  time  difference,  whereas  the 
Dingley  system  utilizes  hyperbolas  derived  from  constant  propagation  frequency  differ- 
ences. 

STANDARD  LORAN.  This  system  operates  at  approximately  2  Mc/s  and  utilizes 
ground  wave  and/or  single  hop  E  layer  paths  of  transmission.  Special  charts  are  provided 
having  Loran  hyperbolas  super- 
imposed on  geographical  maps. 
Figure  7  shows  a  partial  exam- 
ple of  Loran  hyperbolic  charts. 
In  practice,  a  master  (or  A)  sta- 
tion transmits  40-^s  pulses  at 
a  given  repetition  rate.  These 
pulses  serve  to  synchronize  a 
slave  (or  B)  station  several  hun- 
dred miles  away  so  that  the  B 
station  also  transmits  40-/^s 
pulses  at  the  same  repetition  rate, 
but  delayed  in  time  by  the  travel- 
time  between  the  two  stations 
plus  the  B  station's  system  delay. 
The  reception  and  presentation 
of  these  transmitted  pulses 
aboard  ship  by  means  of  a  special 
Loran  receiver  and  indicator 
permit  measurement  of  their 
time  difference,  and  from  that  in- 
formation the  appropriate  Loran 
chart  will  give  the  correct  corre- 
sponding hyperbolic  line  of  posi- 
tion. A  separate  operation  on  a  second  pah*  of  Loran  transmitting  stations  provides  a 
second  position  curve,  and  the  intersection  of  these  hyperbolic  curves  on  the  special  charts 
establishes  a  "fix."  A  block  diagram  of  the  Loran  system  is  shown  in  Fig.  8. 

Transmitting  Equipment.  The  Loran  transmitting  stations  generally  operate  as  a 
group  of  four  stations.  Each  station  would  normally  consist  of  a  timer  and  a  transmitter. 
The  timer  establishes  a  very  precise  pulse  rate  by  utilizing  50  kc/s  or  100  kc/s  crystal- 
controlled  oscillations  of  very  high  accuracy,  which  are  divided  down  to  provide  basic 
pulse  rates  of  25  pps  and  33  Vs  PPS.  These  timing  pulses  trigger  an  exciter  which  generates 
the  40-/ZS  pulses  for  modulating  the  transmitter.  The  transmitter  itself  is  a  self-excited, 
tuned-grid,  tuned-plate,  push-pull  unit,  pulse  modulated  in  the  cathode  circuit  and 
oscillating  from  1750  to  2000  kc/s.  To  provide  precise  synchronization,  to  conserve 
equipment,  and  to  reduce  maintenance,  a  single  transmitter  is  usually  employed  as  a 
master  station  which  synchronizes  two  different  slave  stations,  on  the  same  frequency, 
but  at  different  repetition  rates.  This  is  accomplished  by  providing,  at  the  transmitter, 
two  separate  timers  (each  of  which  is  independently  referenced,  for  convenience,  to  the 
precise  time  signals  from  station  WWV),  two  exciters,  and  a  mixer  stage  preceding  the 
modulator.  The  duty  cycle  of  this  double-pulsed  transmitter  system  is  approximately 
80  fj.s  in  every  40,000  /zs  (25  pps),  or  0.2  per  cent.  The  transmitter,  operating  at  about 
100  kw  peak  power,  feeds  a  120-ft  guyed  radiator  in  the  standard  installation.  In  case  of 
damage  by  windstorm,  etc.,  an  emergency  T  antenna  is  provided.  The  slave-station 


FIG.  7.     Loran  Hyperbolic  Line  of  Position 


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AIDS  TO  NAVIGATION 


emitting  equipment  is  identical  to  that  of  the  master  station,  and  suitable  means  are 
provided  at  the  slave  station  for  properly  receiving  the  master  station  signals  and  for 
effecting  the  necessary  precise  synchronization  of  the  slave-station  pulses. 


Major  components  of 
Loran  receiver 


Major  components  of 

double  pulsed  Loran 

master  transmitting  station 

FIG.  8.     Functional  Block  Diagram  of  Basic  Loran  System 

RECEIVING  EQUIPMENT.  The  craft's  receiver-indicator  equipment  is  usually 
furnished  with  a  50-  to  60-ft  vertical  antenna  leading  to  a  medium-frequency  receiver 
having  a  band  width  of  50  kc/s  to  pass  the  40-Ais  pulses,  and  a  50-kc  video  output  stage. 
To  insure  maximum  range  reception,  the  receiver  must  have  enough  sensitivity  and  gain 
to  provide  full  screen  deflection  with  10  microvolts  into  the  receiver.  The  indicator  pro- 
vides the  necessary  timing  circuits  to  measure  pulse  separations  with  the  required  I-AIS 
precision.  A  100  kc/s  precision  oscillator  with  associated  dividers  supplies  a  sequence  of 
precise  index  timing  markers  at  appropriate  intervals,  with  a  basic  accuracy  of  dbl  us.  It 
also  contains  the  delay  and  deflection  circuits  for  displaying  the  received  pulses  or  the 
timing  markers  on  the  oscilloscope  screen.  The  actual  method  of  presentation  is  to  pro- 
vide separate  paralleled  sweep  traces  on  the  scope  screen,  one  for  the  train  of  A  station 
pulses  and  one  for  the  train  of  B  station  pulses,  with  the  index  markers  appearing  simul- 
taneously on  the  screen  with  the  station  pulses.  Since  more  than  one  Loran  transmitter 
may  be  operating  on  the  same  carrier  frequency,  pulses  from  all  these  stations  will  appear 
on  the  indicator  screen.  Therefore,  the  indicator  includes  timing  and  sweep  circuits  which 
provide  precise  repetition  rates  corresponding  to  the  desired  transmitter's  repetition  rates. 
This  procedure  synchronizes  the  indicator  to  the  pulses  of  only  one  transmitting  pair  of 
Loran  stations  while  the  pulses  of  all  others  continuously  drift  across  the  screen  and  do  not 
interfere  with  the  desired  stationary  pair  of  pulses. 

The  exact  amount  of  time  delay  between  the  slave  and  master  pulses  can  be  measured 
from  manually  incorporated  time  delay  circuits  required  to  make  both  these  pulses  coincide 
on  the  screen.  The  accuracy  of  measurement  depends  on  the  stability  of  the  circuits  which 
provide  timing  markers  on  the  face  of  the  oscilloscope.  To  facilitate  this,  an  expanded 
or  fast  sweep  is  provided,  enlarging  the  picture  of  the  pulses  and  making  the  pulse-matching 
procedure  accurate  to  approximately  I-MS,  since  the  10-jus  marker  spaces  can  be  estimated 
to  tenths.  A  typical  presentation  is  shown  in  Figs.  9  (a)  and  9(6).  A  receiver  refinement 
introduced  by  the  Sperry  Gyroscope  Company  permits  the  local  time  delay  (equivalent 
to  the  time  difference)  to  be  read  directly  in  microseconds  on  a  Veeder-type  counter.  The 
currently  available  shipboard  receiver-indicator  equipment  weighs  between  125  and  235 
Ib,  although  the  corresponding  airborne  equipment  is  a  comparatively  small  unit  weighing 
approximately  25  Ib.  Both  types  consume  200  to  300  watts  from  a  115- volt  power  source. 


RECENTLY  INTRODUCED  NAVIGATIONAL  AIDS       22-51 


One-hop-E 


Performance.  When  used  over  water  Loran  can  provide  navigational  fixes  up  to  750 
nautical  miles  in  the  daytime  and  up  to  1400  nautical  miles  at  night.  The  daytime  range 
is  the  limit  of  ground-wave  propagation  at  the  frequencies  and  power  used.  At  night, 
pronounced  reflection  effects  from  the  E  layer  of  the  ionosphere  provides  the  additional 
range.  Fortunately,  this  ionospheric  reflection  introduces  only  small  errors  which  are 
compensated  by  known  correction  factors  and  allowed  for  in  the  Loran  charts.  When  the 
ship  is  in  a  position  to  receive  both  ground-  and  sky-wave  transmission,  the  indicator 
will  show  two  or  more  sets  of  pulses  where  the  first  set  is  the  ground-wave  pair  and  the 
others  are  various  sky-wave  reflections  as  shown  in  Fig.  9 (a).  Such  reception  does  not 
interfere  with  the  fix  determinations.  The  average  operating  accuracy  of  present  systems 
is  better  than  0.5  per  cent  of  the  distance  measured,  e.g.,  less  than  31/3  miles  in  500  miles, 
but  deteriorates  somewhat  along  the  base  line  of  the  transmitters.  The  accuracy  depends 
on  the  location  of  the  trans- 
mitting stations,  the  stabil- 
ity of  the  timing  circuits, 
and  the  experience  of  the 
operator  in  matching  pulses 
and  estimating  time  be- 
tween 10-/XS  markers. 

Since  the  system  depends 
upon  the  measurement  of 
time  differences  rather  than 
upon  direction  of  propaga- 
tion, it  is  not  subject  to 
the  usual  errors  encountered 
in  direction-finder  systems. 
Factors  affecting  the  propa- 
gation from  a  master  station 
will,  in  general,  also  affect 
the  propagation  from  its 
slave  station.  This  is  espe- 
cially true  as  the  range  in- 
creases, since  the  two  paths 
then  become  approximately 
parallel.  At  the  close  of  the 
war,  the  military  Loran  sys- 
tem comprised  40  trans- 
mitting stations  located  at 
strategic  geographical 
points,  with  an  additional 
10  under  construction; 
these  systems  supplied  in- 


FTG.  9 (a).     Typical  Ground  Wave  and  Sky  Wave  Pulse  Sequence 

from  One  Station.    Signals  to  the  right  of  the  One-hop-E  are  gener- 

ally  disregarded. 


FIG.  9(6).     Basic  Loran  Indicator  Pattern.     A  calibrated  time  differ- 
ence (or  delay)  mechanism  and  an  expanded  sweep  are  used  to  obtain 
1  microsecond  accuracy. 


formation  to  receiver-indicators  installed  on  more  than  3000  surface  vessels.  Among 
the  improvements  which  are  desired  and  may  be  expected  in  the  near  future  are  the  follow- 
ing: (1)  higher-power  transmitting  stations  (about  1000  kw),  (2)  crystal-controlled  oscil- 
lators to  provide  more  stable  carrier  frequencies,  (3)  different  methods  of  station  synchro- 
nization and  pulse-shaping  circuits,  and  (4)  a  totally  automatic  receiver-indicator  system 
requiring  a  minimum  of  technical  operating  skill  and  experience.  These  improvements 
are  desired  in  order  to  overcome  the  present  disadvantages  of:  (1)  limited  range,  especially 
in  the  daytime,  (2)  the  undesired  synchronization  upsets  caused  by  ionospheric  disturb-' 
ances,  (3)  the  need  of  a  specially  trained  operator  for  interpretation  and  maintenance. 
In  addition,  experimental  studies  are  being  conducted  (see  below)  on  LF  Loran  which  is 
an  adaptation  of  standard  Loran  to  the  very  low-frequency  range. 

SS  LORAN.  This  system  (sky-wave  synchronized  Loran)  is  operable  during  the  night 
only  and  is  identical  to  standard  Loran  except  that  master  and  slave  stations  are  sepa- 
rated by  1000  to  1200  nautical  miles  instead  of  the  usual  200  to  300  nautical  miles.  Such 
a  long  base  line  requires  synchronization  of  master  and  slave  stations  by  sky  waves,  since, 
at  the  operating  frequency  of  2  Mc/s,  the  range  of  the  ground-wave  propagation  is  less 
than  the  base  line  itself,  so  that  SS  Loran  can  be  used  only  at  night,  when  sky  waves  are 
strong. 

LOW-FREQUENCY  LORAN.  To  provide  greater  daytime  range,  a  modified  Loran 
system  has  been  designed  which  operates  at  about  180  kc/s  and  offers  the  following  propa- 
gation advantages:  (1)  at  these  low  frequencies  the  propagation  range  of  ground  waves  is 
increased,  especially  over  land;  and  (2)  sky-wave  reflections  become  much  more  stable, 
to' such  a  degree  that  they  can  more  readily  be  interpreted  by  a  navigator.  ™~  **oWi^r 


This  stability 


22-52 


AIDS  TO  NAVIGATION 


results  from  the  fact  that  radio-frequency  energy  at  these  frequencies  does  not  penetrate 
the  E  layer  of  the  ionosphere  and  thus  the  long  trains  of  multiple  night  time  sky  waves, 
present  at  2  Me,  are  absent.  Hence,  the  180  kc/s  sky-wave  receptions  are  much  more 
useful  to  the  operation  of  the  Loran  system  and  it  appears  possible  to  obtain  ranges  of 
1000  or  more  miles  for  satisfactory,  stable  operation  for  day  or  night  over  land  or  sea. 
However,  resolution  (separation)  between  the  ground-wave  and  various  sky-wave  recep- 
tions deteriorates  and  affects  the  accuracy  of  the  system.  At  very  low  frequencies  (below 
180  kc/s),  ground  waves  are  even  less  attenuated,  and  very  reliable  propagation  over 
thousands  of  miles  can  be  expected.  However,  because  very  wide-band  antenna  systems 
are  not  available  at  the  very  low  frequencies  in  view  of  their  prohibitive  size  and  because 
of  current  limitations  of  wide-band  and  pulsed  techniques,  the  pulse  principles  inherently 
required  in  Loran  have  not  yet  been  adapted  to  the  very  low  frequencies  employed.  In 
view  of  these  factors  it  seems  that  100-150  kc/s  is  about  the  lowest  usable  frequency  range 
in  which  pulsed  systems  such  as  Loran  can  be  employed,  with  the  pulse  length  increasing 
to  about  300  jus. 

GEE  SYSTEM.  The  British  Gee  system  of  navigation  operates  in  the  20-85  Mc/s 
frequency  range,  on  the  pulse-propagation-time-difference  principle.  In  nearly  all  respects 
it  is  the  higher-frequency  counterpart  of  the  Loran  system  and  was  independently  de- 
veloped by  the  British.  The  essential  differences  of  the  Gee  system  are:  (1)  higher  operat- 
ing frequency  and  consequent  reduction  of  range ;  (2)  three-slave-station  operation  instead 
of  two;  and  (3)  simultaneous  presentation  of  two  propagation  time  differences  in  order 
that  a  position  fix  can  be  determined  in  one  initial  receiver  operation. 

CONSOL  (BRITISH)  AND  SONNE  (GERMAN).  (See  references  15  and  16.)  Consol 
or  Sonne  is  proposed  as  a  medium-range  navigational  aid  of  the  rotating  beacon  class. 
Radio  navigational  aids  of  this  class  operate  in  much  the  same  manner  as  a  directional 
searchlight  which  revolves  at  a  known  and  constant  rate  with  the  addition  of  a  non- 
directional  visual  light  signal  marking  the  instant  that  the  revolving  light  passes  through 
true  north.  The  observer,  at  the  point  where  the  line  of  bearing  to  the  beacon  is  to  be 
determined,  needs  only  to  employ  a  stop  watch  to  measure  the  time  interval  between  the 
reception  of  the  non-directional  light  signal  and  the  searchlight  signal,  after  which  the 
desired  azimuth  angle  is  computed  from  the  product  of  the  known  angular  rotational 
speed  and  the  time  interval.  In  the  Consol  system  dots  and  dashes  are  put  into  the 
rotating  antenna  pattern  so  that  a  stop  watch  is  not  needed.  The  observer  simply  counts 
the  number  of  dots  and  dashes  between  the  "north"  pulse  and  the  equisignal,  and  obtains 
his  azimuth  from  the  station  thereby.  A  variation  of  this  rotating-beacon  principle 
consists  of  directly  imparting  the  desired  azimuthal  information  to  the  light  beam  proper 
via  correlated  modulation  methods,  e.g.,  by  coding,  intensity  modulation,  or  frequency 
modulation.  Completely  analogous  rotating  radio  beacon  systems  have  also  been  used. 
For  low  and  medium  frequencies,  large  rotating  antennas,  or  fixed  antennas  (e.g.,  Adcock 
type)  with  rotating  goniometer  arrangements  transmitting  a  cosine-law  or  cardioid  pattern, 
have  been  used.  In  other  systems  a  cardioid  or  cosine-law  modulation  pattern  has  been 
transmitted,  utilizing  a  receiver  with  a  pattern  synchronized  cathode-ray  sweep  for  an 
automatic  indicator.  Generally,  radio  frequencies  in  the  VHF  and  UHF  regions  have 
been  favored  because  rotating  antennas  of  reasonable  dimensions  can  be  more  conveniently 

built  to  give  the  desired  sharpness  of  beam 
width  to  the  predominantly  single  lobe  ra- 
diation pattern  usually  employed.  The 
necessarily  high  carrier  frequency  of  these 
systems  unfortunately  limits  them  to  line- 
of-sight  ranges.  Consol  or  Sonne,  on  the 
other  hand,  is  a  rotating-beacon  system 
that  operates  in  the  200-500  kc/s  band 
and  provides  an  accurate  medium-range 
facility.  Use  is  made  of  an  ingeniously 
employed  rotating  multilobe  radiation 
pattern  such  that  sharp  bearing  resolu- 
tions are  realized  at  these  ranges. 

SHORAN.  (See  reference  17.)  The 
Shoran,  or  short  range  navigation,  system 
operates  on  the  principle  of  echo  ranging 
on  two  spaced  beacons  of  the  radar- 
respondor  type.  It  combines  the  accuracy 
of  propagation-time  measurement  with 

FIG.  10.    Functional  Block  Diagram  of  Basic  Shoran    simplicity  of  operation  and  reliability  of 
System  equipments. 


Transmitter 


Ft 

Receiver 

CONTEMPLATED  AND  PROPOSED  NAVIGATIONAL  AIDS      22-53 

Briefly,  Shoran  position  "fixing"  requires  the  measurement  of  the  round-trip  trans- 
mission times,  in  terms  of  range  or  distance,  of  two  sets  of  short  radio-frequency  pulses 
transmitted  by  the  craft,  each  set  interrogating  a  known  shore  radar  respondor-beacon 
station  which  retransmits  the  pulses  back  to  the  craft.  Since  each  echo  range  defines  an 
arc  of  fixed  radius  from  each  known  radar  respondor-beacon  position,  the  intersection  of 
these  arcs  determines  the  craft's  location.  The  resulting  double  ambiguity,  as  in  Loran 
and  Gee,  can  usually  be  easily  resolved.  The  two  sets  of  pulses  are  transmitted  from  the 
craft  on  separate  frequencies  as  shown  in  Pig.  10.  Upon  interrogation,  both  ground 
beacon  stations  retransmit  on  a  common  frequency  that  is  different  from  either  of  the 
.signal  frequencies  radiated  by  the  craft.  The  returned  pulses  are  then  received  and  dis- 
played by  the  craft's  equipment.  The  crystal-controlled  repetition  rate  is  the  precise 
yardstick  in  terms  of  which  the  craft  to  respondor-beacon  distances  are  measured,  and  all 
interrogating  transmitters  use  this  same  repetition  rate. 

DIRECTION-FINDER  SYSTEMS  (HF-VHF-UHF).  (See  references  18,  19,  and  20.) 
Until  recently  direction-finder  systems  at  these  frequencies  were  quite  unreliable.  Pri- 
marily, this  poor  operation  has  been  caused  by  resonant  reradiation  from  structures  near  the 
DF  collector.  Since  a  grounded  metallic  structure  will  resonate  approximately  as  follows: 

/  246  rn 

/(Mc/s)  =  TTT-  (5) 


structures  between  about  8  ft  and  165  ft  high  will  have  a  resonance  somewhere  in  the  1.5  to 
30  Mc/s  frequency  band.  (See  Environmental  Effects,  p.  22-39.)  On  the  basis  of  these 
analyses  shipboard  HF/DF  can  now  be  installed  with  average  errors  of  5-10°. 

In  view  of  their  poor  accuracy  compared  to  the  1°  to  2°  shipboard  MF/DF  accuracy, 
it  is  not  likely  that  shipboard  HF/DF  will  be  widely  employed  for  other  than  military 
intercept  and  location  applications. 

Shore  stations  (see  reference  20)  of  100  Mc/s  direction  finders  employing  a  vertical 
spaced  Adcock  collector,  a  spinning  goniometer,  and  a  cathode-ray-type  automatic  indi- 
cator have  given  a  2-3°  accuracy  with  line-of-sight  range.  At  8  Mc/s  planes  have  been 
tracked  from  the  United  States  to  Northern  Africa.  For  the  ultra  high  frequencies  90  to 
5000  Mc/s  systems  use  a  spinning  reflector-type  collector  (dish)  system  in  lieu  of  the  fixed 
Adcock  and  goniometer  arrangement.  In  general  these  are  not  too  efficient  in  the  90  to, 
say,  200  Mc/s  range  but  rapidly  improve  for  the  higher  frequencies. 

SONAR.  (See  references  21  and  22.)  Sonar,  or  sound  navigation  and  ranging,  which  is 
the  field  of  underwater  sound  navigation,  has  been  applied  in  a  manner  similar,  in  principle 
and  in  electronic  circuits,  to  radar  in  the  field  of  radio  navigation;  the  principal  difference 
is  the  medium  of  propagation.  The  two  main  types  of  underwater  sound  aids  in  use  are: 
(1)  the  vertical  projectors  called  echo  sounding  or  depth  finding  aids;  and  (2)  the  horizontal 
projectors  known  as  echo  ranging  units. 

Echo  sounding  equipment  transmits  sound  pulses  vertically  downward  from  a  projector 
placed  in  the  bottom,  of  the  ship,  or  lowered  from  the  side  of  the  ship,  and  receives  the 
reflected  pulses  from  the  bottom  of  the  ocean  or  waterway,  giving  a  direct  and  constant 
running  record  on  graph  paper  of  the  depth  of  water  beneath  the  boat's  keeL  Present 
commercial  equipment  has  a  range  of  1000  fathoms  or  less  and  is  used  to  navigate  channels, 
rivers,  and  other  waterways  and  to  obtain  data  for  charting  the  depths  of  these  various 
waterways.  Information  necessary  for  the  plotting  of  charts  having  contour  lines  of 
constant  depth  is  gathered  by  the  Coast  and  Geodetic  Survey  and  is  published  by  the 
Navy  Hydrographic  Office  as  an  aid  to  marine  navigation.  This  type  of  equipment  has 
also  been  successfully  used  by  fishing  boats  to  locate  schools  of  fish,  as  they  also  reflect 
sound  pulses. 

Echo  ranging  equipment  transmits  sound  in  a  pattern  which  is  eonically  beamed.  It 
.searches  in  the  horizontal  plane  from  a  projector  which  can  be  rotated  through  360°. 
•Originally,  the  term  Sonar  implied  the  use  of  ranging  equipment  only,  but  it  now  designates 
almost  any  type  of  underwater  sound  equipment  regardless  of  its  use.  This  equipment 
was  originally  developed  for  the  U.  S.  Navy  for  detecting  submarines  and  surface  ships, 
but  it  can  obviously  be  used  in  the  same  manner  for  navigational  and  anticollision  purposes. 
It  presents  the  distance  from  the  ship  to  an  object,  directly  in  yards,  on  a  cathode-ray 
range  indicator,  and  the  bearing  of  the  object  with  respect  to  the  ship's  position  on  a 
bearing  indicator  compass. 

11.  CONTEMPLATED  AND  PROPOSED  NAVIGATIONAL  AIDS 

LANAC.  The  Lanac  or  Zaminar  navigation  and  anticollision  system  is  a  method  of 
.electronic  navigation,  developed  since  the  close  of  the  war,  which  proposes  a  unified  radar 


22-54 


AIDS  TO   NAVIGATION 


1 

1 

1 

I 

1  Transmitter 

I— 

1 

1 
j 

Challenger 

1     Receiver 

1 

1 

1 

,_       _ 

Indicator 

and  identification  navigational  system  for  both  marine  and  air  use  (see  article  6), 
utilizing  a  minimum  of  equipments.  The  Lanac  system  provides  garget  identification, 
coded  radar  beacon,  and  radar  search  operations  from  the  same  basic  equipment.  Essen- 
tially, the  proposed  system  comprises  a  challenger  and  a  replier,  operating  at  approxi- 
mately 1000  Mc/s,  as  shown  in  the  block  diagram  Fig.  11.  The  challenger  can  also  be 
operated  as  a  low-power  search  radar,  for  auxiliary  navigation  and  anticollision  protection. 

The  repliers  are  coded  for  identifica- 
tion and  (in  air  use)  for  altitude  dis- 
crimination. Thus  the  system  pro- 
vides the  essential  advantages  of  radar 
navigation,  but  the  technical  and 
operational  emphasis  is  placed  on  the 
beacon  operation,  which  provides  pos- 
itive point  identification  of  all  stra- 
tegic locations  and  other  ships, 
assuming  that  the  ships  are  properly 
equipped.  Thus,  for  maximum  effi- 
ciency of  operation,  most  other  ships 
and  prominent  geographic  locations 
would  have  to  be  marked  by  respond- 
or-beacons  (repliers).  The  display  of 
beacon  replies  on  a  PPI  scope  as  a 
two-dimensional  polar  plot  of  the  area 
surrounding  the  challenger  is  easier 
and  more  reliably  interpreted  than  a 
corresponding  radar  display. 

A  chart  of  the  marine  services  ren- 
dered by  Lanac  is  shown  in  Fig.  12 
with  the  addition  of  the  auxiliary 
radar  functions  mentioned.  Although 


1 

1 
"I 
| 

Receiver 

i 
1 

r 
i 

Repfler 

1 

"1 

1 

Transmitter 

u 

i 

1 

1 
1 

Identity 
Coding 

i 
J 

FIG.   11.     Functional  Block    Diagram   of   Basic  Lanac 
System 


not  yet  in  general  use,  Lanac  offers  the  advantages  of  a  simple  and  very  comprehensive  nav- 
igational aid  which  combines  the  reliability  of  beacon  operation  with  the  versatility  of  radar. 
DECCA.  (See  reference  16.)  The  British  Decca  system  operates  in  the  low-  and 
medium-frequency  ranges  (i.e.,  10-200  kc/s)  on  the  propagation  phase  difference  principle 
(e.g.,  similar  to  POPI).  Operation  is  accomplished  by  transmitting  CW  signals  from  a 
master  and  a  remote  slave  station  accurately  synchronized  so  that  there  is  set  up  in  space  a 
stable  interference  pattern  of  radiations  having  distinct  loci  of  constant  phase  difference. 

/    Danger-ground 
buoy  beacon 


FIG.  12.     Summary  of  Services  Rendered  by  Lanac  (Excluding  Radar) 

These  Decca  loci  are  hyperbolas  of  constant  propagation  phase  difference,  as  compared 
to  the  Loran  loci  which  are  hyperbolas  of  constant  propagation  time  difference.  To  facil- 
itate the  reception  of  the  two  distinct  transmitted  signals,  and  the  practical  measurement 
of  their  phase  difference,  the  Decca  master  and  slave  CW  emissions  have  a  highly  stabilized 
ratio  of  carrier  frequencies;  e.g.,  the  master  station  may  operate  on  340/4  =  85  kc/s,  the 
slave  station  on  340/3  =  113.33  kc/s.  The  received  frequency  ratio  is  converted  to  unity 
by  suitable  multipliers  in  two  channels  of  the  receiving  equipment.  The  actual  phase 


CONTEMPLATED  AND  PROPOSED  NAVIGATIONAL  AIDS      22-55 

difference  between  the  two  signals  is  then  measured  by  passing  the  two  converted  signals 
through  phase  discriminating  and  indicating  circuits.  An  integrating  phase-meter  (called 
a  decometer)  numerically  identifies  each  hyperbola  of  0°  phase  difference  as  it  is  picked  up 
by  the  receiver.  These  0°  phase  difference  hyperbolas  are  called  "equiphase"  lines  and 
are  used  as  a  reference.  All  other  hyperbolas  within  the  "lanes"  between  the  equiphase 
4  lines  are  then  identified  on  the  decometer  as  values  of  phase  difference  other  than  0°. 
Since  two  such  position  lines  are  required  for  a  "fix,"  another  such  phase  comparison 
determination  is  necessary  on  a  second  pair  of  stations.  The  total  equipment,  therefore, 
comprises  three  shore  transmitters  (one  master  operating  two  slaves)  together  with  their 
associated  control  circuits,  and  craft  equipment  consisting  of  three  phase-stable  amplifiers, 
four  frequency-multipliers,  and  two  integrating  phase-meters. 

In  order  to  avoid  ambiguities  of  fix  in  the  Decca  system,  the  decometers  must  be  preset 
to  a  given  reading  at  a  known  point  which  is  within  reception  range  of  the  Decca  trans- 
mitters. Then,  as  long  as  the  transmitters  and  the  receiving  equipments  are  operating, 
the  decometer  will  read  correctly  as  the  receiver  travels  from  one  lane  to  another.  If 
either  the  receiver  or  any  of  the  transmitters  fail  to  operate  properly,  or  the  receiver  travels 
out  of  reception  range,  the  absolute  setting  is  lost  and  must  be  reset  at  a  known  position. 
To  reduce  these  possibilities  of  ambiguity,  means  for  positive  lane  identification  are  now 
being  developed,  A  Decca  facility  (utilizing  four  stations  centered  near  London,  England) 
is  in  continuous  24-hour  operation.  Other  Decca  facilities  are  being  planned  including 
some  for  the  European  continent.  The  Decca  system  is  easier  to  operate  than  the  Loran 
system  and  appears  capable  of  giving  fix  information  with  greater  accuracy,  but  at  de- 
creased range.  However,  errors  due  to  serious  phase  shifts  may  be  introduced  by  iono- 
sphere reflections  of  sky  wave.  It  is  to  be  noted  that  three  carrier  frequency  channels  are 
required  for  a  position  fix  with,  the  Decca  system. 

POPI.  (See  reference  16.)  The  British  POPI  or  post  office  position  indicator  system, 
operates  in  the  medium-frequency  range  on  the  propagation-phase-differenee  principle 
(similar  to  Decca)  with  such  a  short  base  line  that  the  hyperbolic  lines  of  position  are 
essentially  straight  lines  or  bearing  lines  of  position  to  each  POPI  station.  Two  or  more 
bearing  lines  of  position  from  two  or  more  POPI  stations  give  the  desired  position  fix. 
Each  POPI  station  employs  three  outer  antennas  symmetrically  disposed  about  a  central 
antenna.  If  the  antenna  spacings  are  equal  to  or  less  than  X/2,  there  are  no  ambiguities, 
and  at  a  distance  exceeding  five  times  the  antenna  spacing  the  hyperbolic  lines  may  be 
considered  as  straight  bearing  lines  of  position.  In  operation,  the  central  antenna  transmits 
continuously  at  a  radio  frequency  of /s,  the  outer  antennas  transmit  at  a  slightly  lower  radio 
frequency  /i,  and  they  are  sequentially  keyed  at  some  fraction,  1/n,  of  the  frequency 
difference  /2.  For  a  typical  POPI  station,  /i  may  be  equal  to  750.000  kc/st  /a  =  80  cps, 
/3  =  750.080  kc/s  (obtained  by  mixing  A  with  /2),  and  n  =  4.  With  n  —  4»  the  keying 
cycle  would  occur  as  follows: 

First  outer  antenna  on  first  1/20  sec 
Second  outer  antenna  on  second  1/20  sec 
Third  outer  antenna  on  third  1/20  sec 
Space  and  reference  on  fourth  1/20  sec 

At  the  receiver,  the  audio  beat  frequency  /2  =  /s  —  fi  (e.g.,  80  cps)  is  applied  to  a  rotating 
switch  with  four  contact  sectors.  In  order  to  control  this  rotating  switch  so  that  the 
sequential  transmissions  will  be  correctly  received,  one  of  the  80-cps  outputs  from  the 
rotating  switch  is  used  to  synchronize  an  80-cps  oscillator  which,  in  turn,  drives  a  syn- 
chronous motor.  This  motor,  through  reduction  gearing  proportional  to  the  factor  l/nf 
permits  1/20-Bec  intervals  for  receiving  the  sequential  80-cycle  tones  corresponding  to  the 
frequency  difference  of  the  transmissions  from  the  outer  and  central  antennas  of  the  POPI 
station.  Since  each  of  the  outer  antennas  has  a  separate  distance  or  propagation  time  to 
the  receiver,  there  is  a  correspondingly  separate  and  distinct  phase  for  each  80-cycle  tone 
sector  of  the  synchronous  commutator's  output. 

It  is  planned  to  measure  these  phase  differences  as  if  they  were  coexistent  even  though 
they  actually  occur  in  alternate  time  sequence.  One  means  proposed  for  accomplishing 
this  measurement  consists  of  a  separate  80-cycle  filter  network  for  each^  synchronous 
commutator  segment  which  is  capable  of  ringing  or  sustaining  80-cycle  oscillations  after 
being  energized  for  about  1/20  sec  without  exhibiting  adverse  relative  phase  shifts  between 
networks.  No  report  is  available  at  present  on  results  obtained  with  this  critical  ringing 
network.  The  present  British  developments  are  in  an  experimental  state,  although  full- 
scale  trials  are  contemplated  at  an  early  date.  It  is  to  be  noted  that  POPI  may  suffer 
the  same  errors,  due  to  ionosphere  phase  shifts,  that  are  expected  in  the  Decca  system. 

TELERAN.  (See  reference  16.)  Teleran,  or  ieZevision  radar  and  navigation,  is  a  system 
that  was  originally  proposed  as  an  aid  to  air  navigation  and  is  discussed  in  detail  in  article  6. 


22-56  AIDS  TO  NAVIGATION 

It  is  believed,  however,  that  it  has  considerable  parallel  usefulness  as  a  harbor  traffic 
control  type  of  marine  navigational  aid,  particularly  during  adverse  weather  conditions. 
Basically  Teleran  is  a  television  system  whereby  the  central  control  (shore)  station  trans- 
mits, by  television,  any  essential  information  to  nearby  craft.  In  this  manner  pertinent 
PPI  (plan  position  indication)  information,  weather  warnings,  and  special  instructions 
can  be  instantly  presented  via  the  television  link  to  the  navigator,  using  only  a  standard 
television  receiver  as  the  shipboard  equipment. 

FACSIMILE.  Because  of  the  increased  speed  with  which  navigational  information  will 
be  available  to  the  navigator  in  the  future,  it  is  expected  that  some  storing  and  recording 
means  will  eventually  be  required  in  order  properly  to  handle  navigational  information 
that  cannot  be  too  well  interpreted  via  oscilloscopes,  etc.,  under  busy  harbor  and  related 
conditions.  Moreover,  the  need  and  desire  for  rapidly  printed  records  of  weather,  warn- 
ings, and  congested  traffic  conditions  will  serve  as  a  powerful  stimulus  in  the  eventual 
adoption  of  such  an  aid.  In  view  of  the  relative  ease  with  which  facsimile  signals  can  be 
multiplexed  with  other  transmissions  to  the  craft  being  navigated,  its  introduction  is  not 
expected  to  be  long  delayed.  In  some  respects,  facsimile  transmission  becomes  a  desirable 
compromise  between  the  systems  of  telephoto,  television,  and  teletype  transmissions. 
Details  of  the  principles  of  facsimile  are  discussed  in  Section  19. 

SOFAR.  Sofar,  or  sound  /ixing  and  ranging,  is  a  proposed  system  for  air-sea  rescue 
service  and  is  a  possible  long-range  navigational  aid  in  the  field  of  underwater  sound.  It 
has  been  discovered  that  sound  originating  at  certain  depths  (often  3000  to  4000  ft)  will 
be  refracted  downward  by  the  layers  of  water  above  the  critical  depth,  owing  mainly  to 
the  temperature  gradient  in  the  water,  and  upward  by  the  layers  of  water  below,  owing 
largely  to  the  hydrostatic  pressure  gradient,  thus  confining  the  sound  energy  to  a  hori- 
zontal channel.  It  has  been  found  possible  to  use  this  channel  for  the  transmission  of  a 
distress  signal  (e.g.,  by  detonation  of  a  suitable  explosive  charge)  over  a  range  exceeding 
2500  miles.  Reception  of  this  signal  by  several  coordinated  and  widely  spaced  hydro- 
phone receivers  allows  an  accurate  position  fix  of  the  distressed  craft.  Special  light-weight 
and  properly  armed  charges  have  been  designed  for  airplane  and  life-raft  applications  that 
are  particularly  efficient  for  air-sea  rescue  purposes.  An  initial  network  of  four  stations, 
at  Hilo  and  Kaneoke  in  the  Hawaiian  Islands,  and  at  Monterey  and  Point  Arena  on  the 
California  coast,  principally  for  air-sea  rescue  purposes,  cover  the  long  air  route  from  San 
Francisco  to  Hawaii. 

REDAR.  Redar,  or  red  (and  infrared)  detection  and  ranging,  is  a  proposed  aid  to 
marine  navigation,  similar  in  principle  to  radar.  In  redar,  it  is  proposed  to  use  a  search- 
light and  telescope  arrangement  in  the  near-infrared  band  aboard  the  ship.  The  general 
techniques  and  procedures  employed  in  radar  are  expected  to  be  modified  as  necessary 
and  applied  to  redar.  For  navigational  applications,  where  rapid  redar  scanning  is  not 
necessary,  a  simplified  infrared  optical  range  finder  operating  on  well-known  optical  range 
finding  principles  is  proposed.  At  the  present  state  of  infrared  investigations,  ranges  of 
3  to  5  miles  are  predicted  during  average  to  clear  weather  and  low  humidity  conditions, 
with  accuracies  comparable  to  radar  accuracies.  Because  of  the  nearly  equivalent  attenu- 
ations of  infrared  radiations  as  compared  to  visible  light  radiations  under  fog  conditions, 
infrared  systems  are  not  expected  to  become  a  general  navigation  system.  If  the  need  for 
very  short-range  target  and  obstacle  detection  becomes  pronounced,  it  is  possible  that 
redar  may  adequately  serve  this  need  in  a  manner  superior  to  that  of  most  standard  radar 
systems. 

MISCELLANEOUS  SYSTEMS.  Dingley  system  (see  reference  23)  is  a  frequency- 
modulation  type  of  radio  navigation  aid  that  basically  operates  on  the  pr op agat ion-time- 
difference  principle,  wherein  the  actual  time  difference  is  obtained  in  the  form  of  a  fre- 
q.uency  difference  (or  audio  beat) .  This  is  accomplished  by  the  use  of  frequency-modula- 
tion emissions  from  two  spaced  antennas,  whereby  there  is  set  up  in  space  a  stable  pattern 
of  frequency  differences  having  distinct  loci  of  constant  frequency  difference.  Thus  the 
Dingley  loci  are  hyperbolas  of  constant  frequency  difference  as  compared  to  the  Loran 
loci  or  hyperbolas  of  constant  propagation  time  difference.  In  operation,  a  central  antenna 
and  at  least  one  other  outer  antenna  are  energized  with  frequency-modulation  signals. 
The  signal  received  along  a  radial  line  from  either  antenna  has  a  space  distribution  of 
frequencies  when  that  antenna  is  emitting  frequency /o  at  time  fa  fi  at  time  fr,  fa  at  time  fa 
etc.  At  time  ti  the  signal  of  frequency  /o  has  traveled  to  0*1,  and  signal  of  frequency  /i  is 
being  emitted  at  antenna  A  or  at  do-  The  partial  table  shows  the  overall  conditions  respon- 
sible for  the  space  distribution  of  frequencies.  Since  the  table  is  also  applied  to  give  similar 
space  distribution  of  frequencies  for  similar  emissions  from  the  second  antenna,  it  can  be 
seen  that  those  points  in  space  which  maintain  a  constant  distance  difference  to  the  two 
antennas  will  define  a  hyperbola  along  which  will  be  found  a  constant  frequency  difference. 
Hence,  a  family  of  frequency-difference  hyperbolas  exists  for  the  various  differential  dis- 


OPTIMUM  TRANSMISSION  PARAMETERS  22-57 

tances  to  the  two  antennas,  and  their  general  similarity  to  Loran  hyperbolas  of  constant 
propagation  time  difference  is  immediately  apparent.  If  desired,  the  reference  antenna,  A, 
can  be  designated  as  master  and  any  other  antennas,  B,  C,  etc.,  can  be  designated  as  slaves. 

INSTAN-         INSTAN-  INSTAN-  INSTAN-  INSTAN-  INSTAN- 

TANEOUS         TANEOUS  TANEOT7S  TANEOUS  TAJtfEOUS  TANBOUS 

TIME        FREQUENCY  FBBQUENCT  AT  FREQUENCY  AT  FREQUENCY  AT  FREQUENCY 

AT  do  di  d2  ds  AT  dn 

*o  /o 

*i  /i  /o 

*a  /2  A  /o 

*3  /3  /2  A 

**  /»  /n-1  /n-2  /n-3 

In  view  of  the  above,  the  Dingley  system  may  be  considered  a  simplified  type  of  Loran 
system  operating  on  an  F-M  basis.  Because  of  the  wide  band  factors  generally  associated 
with  frequency  modulators,  it  may  be  expected  that  the  higher  carrier  frequencies  may  be 
required  with  their  correspondingly  reduced  range. 

Sonic  direction  finders  have  long  been  employed  by  the  military  particularly  for  anti- 
aircraft purposes.  Similar  sonic  direction  finders  have  been  employed  for  underwater 
applications.  In  such  applications,  a  pair  of  spaced  sonic  pickup  devices  (e.g.,  crystals 
acting  as  hydrophones  or  sound  microphones)  have  their  outputs  differentially  connected 
to  yield  a  cosine  law  of  coupling  (i.e.,  similar  to  a  pair  of  Adcock  dipoles)  and  have  their 
resultant  output  fed  to  a  suitable  receiver.  When  used  with  a  system  of  underwater  sound 
beacons  the  combined  arrangement  is  similar  to  the  radiobeacon  system  described  under 
Established  Navigational  Aids,  p.  22-36.  Beacon  stations  are  under  investigation  for 
automatically  taking  sonic  bearings  on  a  ship  and  transmitting  by  radio  the  range  and 
direction  of  the  ship  with  respect  to  the  beacon's  position. 

Infrared  systems  have  been  proposed  which  are  essentially  similar  to  redar  with  an 
optional  infrared  beacon  system  on  shore  to  provide  a  convenient  navigational  aid.  The 
infrared  systems,  as  pointed  out  previously,  do  not  permit  navigating  during  fog  conditions 
owing  to  their  inability  to  penetrate  the  fog  without  an  attenuation  almost  as  great  as  that 
experienced  by  visible  light.  Their  chief  value  would  appear  to  be  a  black  light  navigating 
system  operating  under  blackout  conditions  and 'during  the  absence  of  fog.  Such  systems 
generally  operate  in  the  near  infrared  region  (0.3  to  1.5  microns). 

Far  infrared  (heat-ray)  systems  have  been  developed  to  distinguish  targets  from  their 
backgrounds  by  operating  on  heat  or  temperature  differences.  These  systems  generally 
function  in  the  8-  to  15-micron  band  and  usually  operate  in  a  manner  wherein  the  heat 
rays  are  detected  by  bolometers  (see  reference  24).  One  type  of  bolometer  successfully 
developed  by  Johns  Hopkins  University  operates  as  a  superconductor  at  temperatures 
near  absolute  zero.  This  superconducting  device  may  include  scanning  arrangements 
whereby  target  resolution,  or  definition,  roughly  comparable  to  radar  definition,  can  be 
attained.  Because  of  atmospheric  absorption,  ranges  beyond  a  few  miles  are  seldom  ob- 
tained. 

Microwave  thermal  detection  has  been  reported  by  G.  C.  Southworth  (reference  25) 
in  an  interesting  series  of  measurements  of  microwave  radiations  from  the  sun.  This 
phenomenon  is  to  be  expected  since  radio  waves  may  be  considered  as  infrared  radiation 
of  very  long  wavelength,  and  a  hot  body  would  be  expected  to  radiate  microwave  energy 
thermally.  Using  such  techniques,  absorption  bands  of  water  vapor  have  been  measured. 
It  is  to  be  further  expected  that  targets  casting  a  shadow  by  blanking  out  radiations  over 
certain  areas  might  be  detected  by  such  microwave  thermal  and  similar  techniques.  Any 
radiation  source  or  any  radiation  field  arranged  to  cause  target  shadows  is  potentially  a 
means  for  detecting  these  targets,  whether  the  frequency  of  these  radiations  is  sonic  or 
extends  to  the  cosmic-ray  region,  and  whether  or  not  the  emitting  source  is  under  control 
of  the  local  observer. 

12.  DETERMINATION  OF  OPTIMUM  TRANSMISSION  PARAMETERS 
FOR  SOME  LONG-RANGE  RADIO  NAVIGATION  SYSTEMS 

-  BASIC  CONSIDERATIONS.  A  long-ran-ge  transoceanic  radio  navigation  system  for 
global  application  must  meet  three  essential  requirements,  namely,  provide:  (1)  adequate 
signal  reception  with  a  sufficient  number  of  coastal  or  island  stations  for  world-wide 
coverage;  (2)  reliable  signal  reception  at  useful  levels  irrespective  of  weather,  time  of  day, 


22-58  AIDS  TO  NAVIGATION 

season,  year,  direction,  and  distance  of  reception;  and  (3)  an  economically  feasible  arrange- 
ment avoiding  prohibitive  installation  and  operating  costs.  P.  R.  Adams  and  R.  I.  Colin 
(reference  26)  have  studied  these  essential  requirements  and  have  arrived  at  the  conclusion 
that,  if  the  craft  should  at  all  times  be  within  range  of  two  ground  stations  (in  order  to 
obtain  a  cross  fix) ,  then  these  stations  should  have  a  minimum  range  of  1500  miles  in  order 
to  insure  world-wide  double  coverage.  In  addition,  they  have  analyzed  the  relative 
suitability  of  the  various  transmission  parameters  for  a  reliable  1500-mile  range  of  a  long- 
range,  essentially  CW,  radio  navigation  system.  These  analyses  cover  the  optimum  choice  ' 
of  frequency,  band  width,  power,  modulation,  and  radiation  (antenna)  efficiency,  and 
comprehensively  consider  these  transmission  parameters  with  due  respect  to  the  following: 
(1)  Quasi-minimum  field  strength  (defined  as  the  monthly  average  of  signal  strength 
measured  at  the  worst  hour  of  the  day  during  the  worst  months  of  the  worst  year)  that  may 
be  expected  at  different  places,  frequencies,  direction  of  transmissions,  etc.  (2)  Quasi- 
maximum  significant  static  intensity  (defined  as  the  value  of  static  which  is  rarely  exceeded 
at  any  time  and  place  where  the  useful  signal  is  likely  to  have  its  quasi-minimum  field 
strength)  that  may  be  expected  at  different  places  on  different  frequencies,  etc.  (3)  Signal- 
to-noise  ratio,  based  on  the  ratio  of  quasi-minimum  field  strength  to  quasi-maximum 
significant  static  intensity,  in  order  to  determine  the  ratio's  minimum  probable  value,  by 
giving  proper  regard  to  each  factor  in  the  ratio  in  order  to  avoid  their  extreme  values 
when  the  time  and  place  of  their  occurrence  may  never  coincide  (the  time  relationship 
for  example,  depends  upon  factors  such  as  latitude,  time  of  day,  and  direction  of  trans- 
mission) .  (4)  Cost  of  reliability  in  the  choice  of  a  suitable  transmitting  frequency  (aside 
from  the  normal  merit  factors  determining  a  choice  of  transmitting  frequency  based  on 
average  reception  conditions) ,  the  degree  and  frequency  of  occurrence  of  f adeouts,  adverse 
polarization  changes,  static  conditions,  and  susceptibility  to  magnetic  storms.  (5)  Corona 
factors,  since  the  size  of  antenna  is  partly  determined  by  the  required  antenna  capacitance 
that  will  prevent  the  antenna  charging  current  from  developing  a  peak  voltage  in  excess 
of  the  critical  corona  voltage;  the  latter,  in  turn,  is  dependent  upon  the  weather  as  well 
as  the  dimensional  and  electrical  factors. 

As  a  result  of  the  above  types  of  detailed  investigation,  employing  compiled  data  and 
consideration  of  the  pertinent  factors  including  signal  and  static  strength  and  fluctuations, 
and  antenna  efficiency,  the  conclusion  is  reached  that,  in  an  essentially  CW  type  of  trans- 
mitting system  for  navigational  purposes  up  to  a  maximum  distance  of  1500  miles,  ade- 
quate reception  can  be  assured  with  least  power  input  at  a  transmission  frequency  of 
70  kc/s.  These  investigations  show  that  the  reduction  in  time  lost  through  fading  and 
f  adeouts  offsets  the  disadvantage  of  low-frequency  transmission  and  its  associated  lower 
radiation  efficiency  and  higher  static  intensity.  Estimates  of  power  requirements  for 
several  locations  and  antenna  dimensions  for  handling  such  power  without  corona  are  also 
discussed  in  the  above  reference. 

MODULATION  AND  BAND  WIDTH.  The  above  study  makes  little  mention  of 
the  various  methods  of  modulation  by  which  intelligence  may  be  transmitted  at  the  chosen 
carrier  frequency.  Three  paramount  points  must  be  considered:  (1)  the  rate  of  trans- 
mission of  intelligence  on  any  one  frequency  is  directly  proportional  to  the  band  width 
used;  (2)  for  a  normal  uniform  energy  distribution  of  noise  over  the  frequency  spectrum, 
the  amount  of  noise  energy  received  is  directly  proportional  to  the  receiver  band  width; 
and  (3)  it  has  been  operationally  observed  that  present  high-speed  craft  (including  aircraft) 
can  satisfactorily  utilize  navigational  information  furnished  at  a  low  rate  of  intelligence 
reception  (e.g.,  20-cycle  modulation)  by  utilizing  integration  and  average  indication  sys- 
tems. Specifically,  amplitude  modulation  may  vary  from  CW  through  audio  modulation 
to  pulse  modulation  transmission.  CW  transmission  provides  no  intelligence  information 
except  the  direction  of  propagation  of  the  wave;  the  latter  information  is  extracted  by 
the  receiving  station  and  is  not  a  function  of  the  type  of  modulation.  The  receiver  band 
width  required  is  extremely  narrow,  thereby  reducing  band-width  noise  to  the  minimum. 
Receiver-introduced  modulation  methods,  as  described  on  p.  22-42,  may  be  used  to  dis- 
criminate further  against  noise  and  effectively  increase  the  signal-to-noise  ratio  without 
affecting  the  receiver  noise  (which  is  determined  by  the  input  circuits).  Large  bearing 
errors  may  be  experienced  in  pure  CW  systems  at  long  range,  where  sky-wave  transmission 
must  be  used.  These  bearing  errors  cannot  be  detected  easily  but  may  be  reduced  in 
certain  cases  by  means  of  the  receiver  modulation  methods  mentioned.  CW  transmissions 
may  be  identified  by  their  frequency  or  by  means  of  very  slow  keying  or  on-off  schedules 
which  theoretically  require  a  definite  band  width  but  can  be  reduced  to  a  few  cycles,  effec- 
tively approaching  zero. 

AUDIO  MODULATION  can  be  taken  to  include  modulation  frequencies  from  a  few 
cycles  to  several  thousand  cycles  and  does  not  impose  prohibitive  requirements  on  trans- 
mitter or  receiver  design.  However,  in  view  of  the  fact  that  the  rate  of  intelligence  trans- 


OPTIMUM  TRANSMISSION  PARAMETERS  22-59 

mission  need  not  be  high  for  most  navigating  systems,  it  would  seem  preferable  to  use 
band  widths  of  the  order  of  10  to  100  cycles  and  utilize  suitable  receiving  methods  in  order 
to  reduce  receiver  band-width  noise  to  the  minimum.  It  has  been  operationaUy  observed 
that  modulated  radio  signals  have  a  higher  stability  and  accuracy  performance  beyond  the 
ground-wave  propagation  range  than  do  CW  radio  signals.  This  is  apparently  due  to  the 
operator's  ability  to  distinguish  single-path  transmission  from  multiple-path  transmission 
(either^  by  aural  or  visual  methods)  when  modulated  signals  are  employed.  With  CW 
radio  signals  poorer  performance  results  because  it  is  practically  impossible  to  distinguish 
the  more  stable  single-path  ray  from  the  multiple-path  rays  with  their  larger  instabilities 
and  errors.  Improved  signal-to-noise  ratios  on  desired  signals  can  also  be  effected  in  audio 
modulated  transmitting  systems  by  employing  receiver  synchronizing  procedures  to  match 
the  transmitted  modulation  rate,  thereby  integrating  or  averaging  out  random  noise. 
Furthermore,  improvements  in  signal-to-noise  ratios  (e.g.,  bearing  resolution  and  accuracy 
performance)  can  also  be  obtained  in  audio  modulated  systems  by  employing  the  same 
receiver-introduced-modulations  technique  discussed  under  CW  systems  and  described 
on  p.  22-42  under  direction  finders.  Modulated  systems  also  permit  convenient  means  for 
positively  effecting  station  identification  either  by  keying,  time  scheduling,  or  variations 
of  the  modulation  rates.  It  is  important  to  note  that  modulation  rates  lend  themselves 
to  accurate  synchronization  between  transmitters  and  receivers,  not  only  for  improving 
the  signal-to-noise  ratio  but  also  for  providing  a  highly  effective  means  for  efficiently 
employing  the  frequency  spectrum.  The  spectrum  can  be  conserved  by  permitting  a 
number  of  stations  to  operate  on  a  common  carrier  frequency  with  different  known  and 
stabilized  modulation  rates,  allowing  the  receiver  to  select  the  desired  station  by  syn- 
chronizing on  the  proper  modulation.  This  procedure  is  already  in  use  in  various  adapta- 
tions as  in  the  Loran  method  of  synchronized  pulse  modulation,  where  as  many  as  16 
separate  stations  in  a  given  area  may  share  the  common  carrier  frequency  and  are  dis- 
tinguishable at  the  receiver  by  the  different  synchronous  repetition  (i.e.,  modulation)  rates. 
Pulse  modulation  systems  require  a  greater  band  width,  and,  consequently,  the  inherent 
receiver  noise  is  greater.  However,  methods  of  synchronous  pulse  gating  are  being  de- 
veloped which  effectively  reduce  the  noise  and  provide  a  higher  sensitivity  to  pulse  detec- 
tion. The  wide  band  required  by  one  transmitter  employing  pulse  modulation  can  be 
offset  as  in  audio  modulation  by  utilizing  a  number  of  precisely  controlled  repetition  rates 
at  the  same  carrier  frequency  (as  mentioned  above  in  the  case  of  Loran),  and  by  providing 
for  differentiation  between  these  various  rates  at  the  receiver.  The  greatest  single  advan- 
tage of  pulse  modulation  is  the  very  precise  and  highly  convenient  technique  available  for 
time  synchronization,  which  permits  effective  time  differentiation  between  the  various 
transmission  paths  of  the  arriving  wave.  The  improved  signal-to-noise  ratio  advantage 
offered  by  modulation  and  pulse  synchronization  techniques  constitutes  another  great 
advantage.  However,  pulsed  techniques  become  more  difficult  to  apply  as  the  frequency 
decreases. 

The  efficiency  of  pulse  synchronization  techniques  has  been  operationally  proved  in 
several  instances,  such  as  (1)  in  the  India  Theater  during  World  War  II  where  Loran 
operation  was  continuously  maintained  during  the  monsoon  weather,  when  all  communica- 
tion links  were  erratic  or  inoperative;  and  (2)  under  conditions  of  severe  precipitation 
static  in  Greenland  when,  again,  Loran  provided  the  only  operative  navigation  link, 
whereas  all  communication  circuits  became  unintelligible.  The  advantages  of  pulse 
techniques  for  maintaining  continuous  and  reliable  service  must,  therefore,  be  thoroughly 
investigated  and  included  in  any  comprehensive  analysis  of  the  final  choice  of  the  operating 
parameters  for  a  long-range  world-wide  navigation  system. 

Other  modulations  in  addition  to  the  various  types  of  amplitude  modulation  exist,  such 
as  frequency  modulation,  phase  modulation,  and  pulse-time  modulation.  Although  no 
extensive  study  has  been  made  of  these  for  long-range  navigational  purposes  it  is  well 
known  that  frequency  and  phase  modulation  possess  considerable  noise-reducing  properties 
in  addition  to  the  modulation  synchronization  techniques  previously  described  for  improv- 
ing signal-to-noise  ratios.  However,  these  advantages  are  partly  offset  by  their  usual 
wide-band  transmission  requirements  and  their  general  limitation  to  the  higher  frequencies 
(in  order  to  achieve  adequate  depth  of  modulation)  where  line-of-sight  range  restrictions 
prevail.  Pulse-time  modulation  (PTM)  offers  the  advantage  of  increased  intelligence  on  a 
pulse  channel  but  decreases  the  number  of  simultaneous  channels  allowable  at  one  fre- 
quency. It  also  makes  the  separation  of  single  and  multiple  path  transmissions  more  dif- 
ficult and  prevents  use  of  the  precise  synchronization  techniques  necessary  for  effective 
improvements  in  signal-to-noise  ratios.  Lastly,  hybrid  or  combination  types  of  modula- 
tion and  second  and  third  derivative  frequency  modulation  may  play  increasingly  impor- 
tant roles  as  investigation  continues  into  the  techniques  of  such  systems. 


22-60  AIDS  TO  NAVIGATION 


BIBLIOGRAPHY 

1.  Comdr.  Benjamin  Dutton,  Navigational  Nautical  Astronomy.     TJ.  S.  Naval  Institute  (1942). 

2.  Aids  to  Marine  Navigation.     TJ.  S.  Coast  Guard  (1940). 

3.  Marine  Radiobeacons,  Engineering  Instructions,  Chapter  41.     U.  S.  Coast  Guard  (1944). 

4.  R.  Keen,  Wireless  Direction  Finding.     Iliffe  &  Sons,  London  (1938). 

5.  M.  K  Goldstein,  Naval  Research  Laboratory,  R-1896  (HF/DF  aboard  the  U.S.S.  CORRY-DD463); 

R-1938  (Specifications  for  Minimum  DF  Site  Requirements);  R-2229  (Interpolation  Charts  for 
HF/DF  Shipboard  Calibrations);  R-2469  (HF/DF  Collector  Locations  on  a  PV-1  Navy  Patrol 
Bomber).  Available  through  the  U.  S.  Department  of  Commerce,  Washington,  D.  C. 

6.  M.  K.  Goldstein,  Naval  Research  Laboratory,  Direction  Finder  patent  applications,  Serial  Nos. 

593,900  (Filed:  5/15/45,  Navy  Case  5280)  and  443,899  (Filed:  5/5/42,  Navy  Case  3176);  and 
Patent  2,405,203  issued  8/6/46. 

7.  Fundamentals  of  Radar,  Wireless  World,  October  1945,  pp.  299-302;  November  1945,  pp.  326-329; 

December  1945,  pp.  363-365;  January  1946,  pp.  23-26;  February  1946,  pp.  55-56. 

8.  E.  G.  Schneider,  Radar,  Proc.  I.R.E.,  Vol.  34,  No.  8,  528-578  (August  1946). 

9.  John  H.  DeWitt,  Jr.,  Technical  and  Tactical  Features  of  Radar,  J.  Franklin  Inst.,  February  1946, 

pp.  97-123. 

10.  Principles  of  Radar.     M.  I.  T.  Radar  School  (1944). 

11.  Electronic  Navigation  Aids.     U.  S.  Coast  Guard,  Public  Information  Division  (April  1946). 

12.  Minimum  Specification  Brief  Nos.  1,  #,  and  3  Submitted  as  a  Guide  for  Voluntary  Use  by  Parties 

Interested  in  Navigational  Radar.    U.  S.  Coast  Guard  (Nov.  14,  1945,  revised  Aug.  1,  1946). 

13.  J.  A.  Pierce,  An  Introduction  to  Loran,  Proc.  I.R.E.,  Vol.  34,  216-234  (May  1946). 

14.  The  Loran  System,  Electronics,  Vol.  18  (November  1945) ;  Vol.  18  (December  1945) ;  Vol.  19  (March 

1946). 

15.  John  E.  Clegg,  Consol,  Beacon  Direction  Finding  Systems  of  High  Accuracy,  Wireless  Worldt 

July  1946,  pp.  233-235. 

16.  Report  of  Electronic  Subdivision  Advisory  Group  on  Air  Navigation,  Bulletin  200,  ATSC  Eng. 

Div.  (February  1946).    Available  through  TJ.  S.  Army  Air  Forces,  Wright  Field,  Dayton,  Ohio. 

17.  A.  Seeley,  Shoran  Precision  Radar,  Electrical  Engineering  (Transactions),  April  1946,  pp.  232-240. 

18.  M.  K.  Goldstein,  Proximity  Effect  of  Metallic  Fences  on  Direction  Finders,  Naval  Research  Lab- 

oratory Report  R-1890. 

19.  Spinning  Loop  Type  Direction  Finders  proposed  by  M.  K.  Goldstein,  1942,  see  Naval  Research 

Laboratory  Report  R-284?. 

20.  M.  K.  Goldstein,  Facility  and  Installation  Requirements  for  DAJ/DF  Coastal  Networks,  Naval 

Research  Laboratory  Report  R-8079. 

21.  Early  Echo  Ranging  Sonar,  Electronics,  Vol.  19,  274-282  (July  1946);  Echo  Ranging  Sonar,  Elec- 

tronics, Vol.  19,  88-93  (August  1946). 

22.  Lt.  Comdr.  D.  H.  Macmillian,  RNR,  Precision  Echo  Sounding  and  Surveying.     Henry  Hughes  & 

Son,  London  (1939). 

23.  E.  N.  Dingley,  Jr.,  A  True  Omnidirectional  Radio  Beacon,  Communications,  Vol.  20,  5,  6,  and  35 

(January  1940). 

24.  Papers  on  Detection  of  Infra-Red  Radiations  Given  at  Winter  Meeting  of  The  Optical  Society  of 

America  held  March  1946.  Abstracted  in  Journal  of  the  Optical  Society  of  America,  Vol.  36, 
No.  6  (June  1946). 

25.  G.  C.  Southworth,  Microwave  Radiation  from  the  Sun,  J.  Franklin  Inst.,  Vol.  239,  285-297  (April 

1945). 

26.  Frequency  Power  and  Modulation  for  a  Long-Range  Radio  Navigation  System,  Electrical  Commun- 

ication, Vol.  23,  144-158  (June  1946). 


SECTION  23 
MEDICAL  APPLICATIONS  OF  ELECTRICITY 


BY 

CHARLES  WEYL 
S.  REID  WARREN,  JR. 


ELECTROTHERAPY  AND  SHOCK  THERAPY 

AHT.  PAGE 

1.  Apparatus 02 

2.  Electrotherapy 03 

3.  Shock  Therapy 04 

DIATHERMY  AND  HIGH-FREQUENCY 
SURGERY 

4.  Apparatus 04 

5.  Diathermy  Technique 06 

6.  High-frequency  Surgery 06 

THE  MEDICAL  USES  OF  ULTRAVIOLET 
AND  INFRARED  RADIATIONS 

7.  Apparatus 06 

8.  Therapeutic  Use  of  Ultraviolet  Radiation     07 

9.  Therapeutic  Use  of  Infrared  Radiation. .     08 

ELECTROCARDIOGRAPHY  AND  ELECTRO- 
ENCEPHALOGRAPHY 

10.  Apparatus 08 

11.  Techniques 10 


ELECTROACOUSTIC  DEVICES 

ART.  PAGE 

12.  Aids  to  the  Deaf 11 

13.  The  Stethophone 11 


ROENTGEN  THERAPY 

14.  Purpose  and  General  Technical  Require- 

ments.       12 

15.  Technique 13 


ROENTGENOGRAPHY  AND  ROENTGEN- 
OSCOPY 

16.  Purpose,    General    Technical    Require- 

ments, and  Technique.  . . , 14 

17.  Apparatus 15 

18.  Miscellaneous  Devices 16 

HIGH-VOLTAGE  SHOCK  AND  X-RAY  BURN 

19.  High-voltage  Shock 17 

UO.  X-ray  Burn 18 


23-01 


MEDICAL  APPLICATIONS  OF  ELECTRICITY 

By  Charles  Weyl  and  S.  Reid  Warren,  Jr. 

During  the  latter  part  of  the  eighteenth  century  Galvani  and  Volta  discovered  that  the 
muscles  of  dead  frogs'  legs  could  be  made  to  move  by  contact  with  metals.  The  descrip- 
tions of  their  experiments  indicated  that  they  were  establishing  a  difference  of  electric 
potential  between  nerves  and/or  muscles  of  the  frogs'  legs  and  that  this  applied  difference 
of  potential  caused  a  motion  of  the  muscles.  Since  that  time,  many  techniques  have  been 
devised  for  using  electrical  apparatus  for  studying  physiological  functions,  for  the  diag- 
nosis of  disease,  and  for  therapeusis.  During  the  twentieth  century  the  development  of 
these  medical  applications  of  electricity  has  become  accelerated  and,  particularly  since 
1925,  new  methods  have  been  developed,  tested,  and  applied  to  clinical  practice.  Al- 
though some  of  these  techniques  have  become  standardized  by  accurate  experimental  in- 
vestigations, nevertheless,  the  biophysical  explanations  of  many  of  the  techniques  are  al- 
most completely  unknown.  Although  frequently  the  electrical  apparatus  and  the  tech- 
niques for  their  use  are  simple,  an  accurate  knowledge  of  the  results  of  the  technique  has 
been  impeded  by  differences  in  individual  patients,  by  lack  of  knowledge  of  the  physio- 
logical factors  involved,  and  by  a  lack  of  proper  control  of  the  apparatus  and  technique. 
Thus  there  exists  a  fertile  field  for  investigation  in  which  physicists  and  engineers  are 
aiding  physicians  in  problems  of  electromedical  diagnosis  and  therapeusis. 

With  the  accelerated  development  of  the  electromedical  methods  which  were  known 
before  1925,  and  also  of  the  new  methods  which  have  been  introduced  since  that  time,  there 
has  been  a  great  increase  in  the  literature  of  the  field.  Furthermore,  the  applications  of 
many  of  the  newer  devices  and  techniques  to  medicine  are  subsidiary  to  their  applications 
in  other  scientific  fields.  This  section  is  organized  on  the  basis  of  these  factors.  Each 
article  contains  the  following  items:  (1)  a  list  of  the  fundamental  components  of  the  elec- 
trical apparatus  required  for  a  particular  electromedical  technique  and  an  example  of  their 
combination  into  a  practical,  useful  apparatus;  (2)  a  brief  description  of  the  electromedical 
technique  and  examples  W  its  use;  (3)  a  list  of  references  to  papers  and  books  in  which  the 
electromedical  apparatus  and  technique  are  discussed  in  detail. 

The  literature  of  the  medical  applications  of  electricity  uses  terms  quite  different  from 
those  employed  by  physicists  and  engineers,  and  some  of  them  are  extremely  confusing. 
For  example,  the  term  electrotherapy  is  sometimes  used  as  a  generic  to  represent  nearly  all 
the  electromedical  techniques  except  those  involving  x-rays;  it  also  often  represents  solely 
those  techniques  described  in  the  next  article  involving  the  conduction  through  parts  of 
the  body  of  direct,  pulsating,  and  alternating  currents. 


ELECTROTHERAPY  AND  SHOCK  THERAPY 

1.  APPARATUS 

The  apparatus  employed  for  generating  direct  and  alternating  currents  for  electro- 
therapy has  many  forms,  all  of  which  are  fundamentally  of  simple  design  from  the  engi- 
neering point  of  view.  The  requirements  are  as  follows:  (1)  direct  current,  0  to  80  volts, 
0  to  50  ma,  with  a  continuous  resistance  control;  (2)  alternating  current  at  commercial 
frequencies  from  0  to  50  volts  rms,  0  to  25  ma;  (3)  a  method  for  periodically  surging  or 
modulating  the  alternating  currents  mentioned  in  (2) ;  (4)  an  induction  coil  operated  by 
direct  current  with  a  mechanical  interrupter;  (5)  a  device  for  providing  pulsating  current 
at  frequencies  from  10  to  100  pulses  per  second. 

Occasionally  an  attempt  is  made  to  construct  an  apparatus  which  will  supply  all  these 
wave  forms.  Usually,  however,  the  physician  purchases  different  pieces  of  apparatus  for 
each  purpose.  At  present,  the  trend  is  toward  the  construction  of  apparatus  which 
operates  from  an  a-c  source  of  110  volts,  60  cycles.  For  example,  the  d-c  generator  may 
take  the  form  of  a  small  1 : 1  ratio  transformer  with  the  secondary  connected  to  a  full-wave 
rectifier  and  filter  system.  Across  the  filter  system  is  connected  a  potentiometer,  and  the 
electrodes  are  connected  to  two  terminals  of  the  potentiometer.  The  surging  devices  are 
generally  motor-driven  variable  resistances.  The  apparatus  used  for  these  purposes  has 
not  been  standardized  and  therefore  it  is  not  possible  to  give  typical  wiring  diagrams. 

23-02 


ELECTROTHERAPY  23-03 

The  arrangements  for  protection  of  the  patient  from  excessive  electric  shock  are,  in  many 
installations,  inadequate. 

A  number  of  accessory  parts  comprising,  for  example,  specially  designed  electrodes  and 
a  timer  for  controlling  the  length  of  treatment  are  required. 

The  development  of  technique  in  electrotherapy  is  based  entirely  on  empirical  results 
frequently  acquired  from  doubtful  interpretations  of  meager  data.  Since  remarkably 
little  attention  has  been  paid  to  the  precise  measurement  of  voltage  and  current,  many  of 
the  accumulated  data  are  of  doubtful  value. 

^  Treatments  usually  have  a  duration  of  1  to  15  xnin.  In  general  the  physician  specifies 
his  technique  by  a  measurement  of  the  current  which  flows  through  the  patient,  and  the 
duration  of  the  treatment. 

2.  ELECTROTHERAPY 

It  has  been  known  for  many  years  that  passage  of  electric  currents  through  the  human 
body  may  cause  important  physical  and  physiochemical  changes  within  the  tissue.  The 
following  results  form  the  basis  of  methods  of  diagnosis  and  treatment  by  the  passage  of 
electrical  currents  through  the  body:  (1)  the  transfer  of  ions  from  outside  the  body  into 
the  treated  part;  (2)  the  transfer  of  ions  from  within  the  body  to  an  electrode  outside;  (3) 
the  production  of  heat  within  the  living  tissue;  (4)  the  stimulation  of  nerve  and  muscle 
fibers. 

To  accomplish  these  purposes,  direct  currents,  pulsating  currents,  and  alternating  cur- 
rents are  used.  The  following  paragraphs  describe  the  applications  of  each  of  these  sev- 
eral forms  of  current. 

ELECTROCHEMICAL  CAUTERIZATION.  A  flow  of  direct  current  (called  galvanic 
current  in  electrotherapy)  through  the  body  generally  serves  its  therapeutic  purpose  by 
destroying  diseased  tissue.  If  platinum  electrodes  are  applied  to  a  diseased  part  which  is 
near  the  surface  of  the  body,  and  a  direct  current  is  made  to  flow  between  the  electrodes, 
an  acid  tends  to  form  by  electrolytic  action  at  the  anode,  and  an  alkali  at  the  cathode. 
The  destructive  action  of  such  chemicals  is  well  known,  and  frequently  it  is  possible  to 
control  the  destruction  of  tissue  by  means  of  the  electrical  method  for  the  production  of 
these  chemicals.  Such  a  procedure  is  called  electrochemical  cauterization.  Usually  only- 
one  metallic  electrode  is  placed  in  contact  with  the  tissue.  The  other  electrode,  called  the 
indifferent  electrode,  consists  of  a  metal  disk  several  inches  in  diameter  held  against  a  pad 
of  cotton  soaked  in  a  solution  of  sodium  chloride  and  placed  against  the  skin  of  the  pa- 
tient. The  transfer  of  ions  from  the  salt  solution  apparently  has  negligible  effect  on  the 
human  body.  Electrochemical  cauterization  is  used  for  removing  superfluous  hair,  warts, 
and  small  rodent  ulcers,  and  also  in  the  treatment  of  certain  skin  diseases. 

IONTOPHORESIS.  If  one  hand  of  the  patient  is  placed  hi  a  liquid  electrolyte,  and  this 
electrolyte  is  used  as  an  electrode,  it  is  possible  by  the  application  of  direct  current  to> 
transfer  ions  from  the  solution  into  the  body.  This  process  is  called  iontophoresis.  The 
physiological  results  of  treatment  by  this  method  have  not  been  adequately  explained. 
Some  work  has  been  done  for  the  purpose  of  effecting  local  anesthesia  by  the  transfer  of 
ions  from  cocaine  into  the  body  at  chosen  places.  Metallic  ions  have  been  introduced, 
and  it  is  claimed  that  the  recombinations  that  occur  within  the  body  provide  small  quan- 
tities of  atomic  metal  or  of  metallic  salts  at  the  site  of  disease.  Other  chemicals  such  as 
acetylbetamethylcholine  chloride  and  histamine  have  been  used  for  the  treatment  of 
specific  disorders. 

CATAPHORESIS.  If  a  semi-solid  electrolytic  gelatin  is  formed  in  the  shape  of  a 
cylinder  and  two  metal  plates  are  placed  at  the  ends  of  the  cylinder,  a  direct  current  may 
be  made  to  flow  through  the  gelatin.  After  a  few  minutes  have  elapsed,  there  is  an  in- 
crease of  water  at  the  cathode  and  a  decrease  of  water  at  the  anode.  This  phenomenon  is 
apparently  caused  by  changes  in  osmotic  pressure  due  in  turn  to  the  current  flow.  This 
procedure  is  sometimes  used  to  remove  undesired  liquids  from  skin  lesions.  It  is  called 
cataphoresis. 

MISCELLANEOUS.  It  has  been  found  that  muscles  which  are  in  a  state  of  fatigue 
become  considerably  strengthened  after  treatment  with  direct  current.  It  has  also  been 
claimed  by  many  workers  that  the  passage  of  current  causes  exhilaration  of  some  patients, 
and,  on  the  other  hand,  with  a  slightly  different  technical  procedure,  the  same  type  of 
treatment  may  produce  drowsiness.  The  causes  of  these  effects  are  unknown. 

The  passage  of  small  direct  currents  through  the  body  causes  no  pain.  However,  a 
sudden  change  in  the  amplitude  of  the  current  produces  muscular  twitching  and  pain. 
Normally,  d-c  treatments  require  a  source  capable  of  adjustment  from  0  to  SO  volts.  The 
current  passing  through  the  body  may  be  from  1/2  to  50  ma.  Because  of  the  muscular 
twitching  described  above,  it  is  particularly  important  that  the  current  be  adjusted  slowly 


23-04  MEDICAL  APPLICATIONS  OF  ELECTRICITY 

to  the  desired  value.  Quantitatively,  it  is  usual  to  increase  the  current  linearly  with  time 
at  the  rate  of  1  ma  per  min.  The  d-c  resistance  of  the  two  arms  and  the  trunk  of  the 
average  adult  is  approximately  1400  ohms.  This  resistance  is  measured  by  suspending 
the  hands  in  salt  solutions,  thereby  overcoming  the  high  resistance  of  dry  skins. 

The  passage  of  alternating  current  of  frequency  less  than  a  few  thousand  cycles  per 
second  involves  a  sensation  of  pain  and  twitching  of  the  muscles.  The  intensity  of  mus- 
cular reaction  and  pain  increases  with  the  amplitude  of  the  current  and  with  the  frequency 
up  to  an  indefinite  limiting  value  of  several  thousand  cycles  per  second.  When  the  fre- 
quency is  raised  above  this  value,  the  sensation  decreases  and  finally  disappears  unless  the 
amplitude  of  the  current  is  sufficient  to  produce  appreciable  heat. 

The  steady  application  of  alternating  currents  of  a  few  milliamperes  of  commercial 
frequencies  is  extremely  painful  and  dangerous.  Therefore  a  device  is  required  to  produce 
a  surging  sinusoidal  current  which  may  be  described  in  the  following  manner.  At  the 
beginning  of  the  treatment  the  alternating  current  is  zero  amplitude.  Over  a  period  of 
the  order  of  2  sec  it  is  increased  to  a  maximum  and  decreased  to  zero.  There  follows  a 
rest  period  of  about  2  sec,  and  then  the  surge  is  repeated.  Apparently  the  application  of 
60-cycle  alternating  currents  causes  contraction  and  extension  of  the  muscles.  Treat- 
ment by  this  method  has  been  recommended  for  paralyzed  muscles  or  muscles  damaged  by 
disease. 

FARADIC  AND  MISCELLANEOUS  WAVE  FORMS.  Various  other  types  of  alter- 
nating and  pulsating  currents  have  been  recommended  for  therapeutic  use.  It  will  suffice 
to  list  a  few  of  these:  (1)  the  secondary  current  from  an  induction  coil,  called  faradic  cur- 
rent in  electrotherapy;  (2)  currents  produced  by  the  periodic  charge  and  discharge  of  con- 
densers; (3)  pulsating  current  of  various  wave  forms  produced  by  mechanical  interrupters; 
(4)  high-frequency  brush  discharge  by  means  of  a  Tesla  coil.  Standardization  of  such 
methods  will  require  a  careful  oscillographic  analysis  of  the  electrical  parameters  combined 
with  a  statistical  analysis  of  the  clinical  results. 

3.  SHOCK  THERAPY 

In  certain  mental  disorders  it  has  been  found  that  if  a  convulsive  shock  is  produced  in 
the  patient  by  means  of  drugs,  or  by  electric  currents  passing  through  the  brain,  improve- 
ment in  the  condition  of  the  patient  is  often  observed.  The  techniques  of  shock  therapy 
are  still  based  chiefly  on  empirical  data.  It  is  to  be  noted  that  the  threshold  at  which 
electric  shock  of  the  brain  will  cause  convulsion  is  relatively  close  to  the  higher  lethal 
threshold.  It  is,  therefore,  necessary  to  control  the  shock  precisely.  Shock  treatments 
are  usually  given  in  a  series  of  from  six  to  twelve  treatments  during  periods  of  a  few  weeks. 

It  is  found  that,  using  electrodes  placed  approximately  over  the  two  temples,  currents 
of  the  order  of  1  amp,  adequate  to  produce  convulsion,  are  caused  to  now  by  a  60-cycle 
alternating  voltage  of  approximately  100  volts.  Thus,  the  apparatus  for  shock  therapy 
consists  essentially  of  a  transformer,  the  output  of  which  can  be  continuously  varied,  and 
a  timer  capable  of  producing  exposures  from  0.1  sec  to  0.6  sec  in  steps  of  0.1  sec.  Motor- 
driven  timers,  time-delay  relays,  and  electronic  timers  have  been  used  to  produce  the  de- 
sired intervals  of  shock.  In  order  to  avoid  fatal  accidents,  it  is  important  in  these  devices 
to  use  circuits  such  that  the  failure  of  a  component  will  result  in  a  great  decrease  in  the 
output  voltage. 

BIBLIOGRAPHY 

Cumberbatch,  E.  P.,  Essentials  of  Medical  Electricity.     Mosby  (1929). 

Glasser,  Otto,  Editor,  Medical  Physics.     Year  Book  Publications,  Inc.,  Chicago,  IU.  (1944).     Physical 

therapy,  low-frequency  currents,  pp.  1068-1073. 

Kovacs,  Richard,  M.  D.,  Electro-therapy  and  the  Elements  of  Light  Therapy.  Lea  and  Febiger  (1932). 
Osbourne,  S.  L.,  and  H.  J.  Holmquest,  Technic  of  Electro-therapy  and  Its  Physical  and  Physiological 

Bases.     Charles  C.  Thomas,  Springfield,  IU.  (1944). 
Electric  Convulsion  Therapy  in  Mental  Disorders,  Psychiatric  Quart.,  Vol.  14,  719  (1940). 

DIATHERMY  AND  HIGH-FREQUENCY  SURGERY 

4.  APPARATUS 

Generators  of  high-frequency  alternating  current,  from  750  kc  per  sec  to  3000  Me  per 
sec,  have  been  used  to  raise  the  temperature  of  parts  of  the  human  body  in  the  techniques 
known  as  diathermy  and  high-frequency  surgery. 


DIATHERMY   AND   HIGH-FREQUENCY   SURGERY      23-05 


There  are  two  general  types  of  diathermy  apparatus.  In  the  first  type,  a  high-voltage 
transformer  is  connected  to  an  oscillatory  discharge  circuit  consisting  of  a  spark  gap,  con- 
densers, and  an  inductance.  Loosely  coupled  to  this  oscillatory  circuit  is  a  secondary 
circuit  to  which  the  patient  is  connected.  A  high-frequency  ammeter  is  connected  in 
series  with  the  patient  to  measure  the  magnitude  of  the  treatment  current.  Figure  1  is  a 
diagram  of  such  a  generator.  More  recently  vacuum-tube  oscillators  (see  Section  7)  have 


CD  II® 

FIG.  1.    Diathermy  Apparatus  Using  Spark-gap  High-frequency  Generator 

1,  Step-up  transformer;  2,  spark  gaps;  3,  condenser,  and£4/ inductance  forming  oscillatory  circuit 
which  resonates  at  500-1000  kc;  5,  coil  coupled  to  oscillatory  circuit;  6,  7,  condensers;  8,  high- 
frequency  a-c  ammeter;  9,  leads  to  electrodes. 

been  designed  for  this  purpose.  They  are  connected  in  a  manner  similar  to  that  shown  in 
Fig.  2.  It  will  be  noted  that  the  two  circuits  are  identical  in  all  essential  features  with  the 
two  types  of  radio  transmitters  known  respectively  as  spark  and  continuous-wave  trans- 
mitters. 

Various  techniques  are  used  for  producing  a  high-frequency  electromagnetic  field  within 
the  part  that  is  to  be  treated.  Thus,  the  output  of  the  oscillator  is  sometimes  applied  to 
two  metallic  plates  between  which  the  part  of  the  body  to  be  treated  is  placed.  Alter- 
natively the  output  of  the  oscillator  may  be  fed  to  a  cable  which  can  be  wrapped  in  the 


FIG.  2.     Diathermy  Apparatus  Using  Vacuum-tube  Oscillator 

1,  110-volt,  60-cycle  a-c  supply;  2,  plate  voltage  transformer;  3,  vacuum  tube;  4,  5,  oscillating  cir- 
cuit; 6,  grid  leak  and  grid  condenser;  7,  blocking  condenser;  8,  r-f  choke;  9,  inductively  coupled  output 
coil;  10,  output  leads  to  ammeter  and  patient.  The  cathodes  of  the!  vacuum  tubes  are  heated  by  cur- 
rent from  an  additional  secondary  coil  (not  shown)  of  transformer  2. 

form  of  a  solenoid  around  a  body  part  or  shaped  into  a  spiral  coil  (the  technical  term^is 
pancake)  which  can  then  be  fixed  against  one  side  of  the  part  to  be  treated.  The  special 
electrodes  required  for  surgery  are  described  in  article  6. 

Apparatus  using  a  magnetron  operating  at  2450  Me  is  now  available.  Radiation  ia 
emitted  by  one  of  several  parabolic  reflectors  6  to  10  cm  in  diameter  excited  by  a  dipole. 
The  radiation  is  thus  confined  to  a  relatively  sharp  beam. 

Because  of  the  very  important  danger  of  interference  with  communication  facilities, 
the  F.C.C.  has  prescribed  the  frequencies  permitted  for  diathermy  and  the  frequency 


23-06  MEDICAL  APPLICATIONS  OF  ELECTRICITY 

stability  that  is  required.  The  frequencies  so  assigned  are:  13,560  kc  =fc  6.78  kc; 
27,120  kc±  160.00  kc;  40,680  kc  ±  20.00  kc;  2450  Me  ±  50  Me,  available  for  experi- 
mental work. 

5.  DIATHERMY  TECHNIQUE 

If  the  frequency  of  the  alternating  current  applied  to  the  human  body  is  indefinitely  in- 
creased, an  ill-defined  point  can  be  found  (order  of  magnitude  of  10,000  cycles  per  second) 
beyond  which  no  sensation  other  than  warmth  is  felt.  The  minimum  amplitude  of  alter- 
nating current  which  produces  pain  by  passage  through  the  human  body  varies  with  the 
frequency  of  the  alternating  current  and  with  the  subject.  Average  values  obtained  by 
a  number  of  workers  are  as  follows: 

AVERAGE 

FREQUENCY,  TOLERANCE, 

cycles  per  second  milliamperes 

60  3-    8 

10,000  tolerance  current,  increasing  gradually  to 25-  30 

100,000  tolerance  current,  increasing  quickly  to 250-600 

Heating  effects,  only,  are  noted  above  100,000  cycles  per  second.  As  a  result  it  is  pos- 
sible to  increase  the  current  flowing  through  the  body  to  values  as  high  as  5  amp.  This 
causes  a  sensible  dissipation  of  heat  within  the  tissue.  For  this  purpose  frequencies 
-above  750  kc  per  sec  are  used.  It  has  been  discovered  that  heat  produced  internally 
is  effectual  in  the  treatment  of  certain  diseases.  The  name  diathermy  has  been  given 
to  the  treatments  characterized  by  the  internal  heating  effect  of  high-frequency  alter- 
nating currents.  Electrodes  several  square  centimeters  in  area  are  used  in  order  that 
the  current  may  be  distributed  approximately  uniformly  over  a  large  area.  Other  ma- 
chines in  which  the  patient  is  not  in  contact  with  electrodes,  but  is  placed  in  a  strong  high- 
frequency  electromagnetic  field  (between  "condenser"  electrodes),  have  been  constructed 
for  the  purpose  of  heating  the  entire  body  or,  more  commonly,  large  portions  of  the  body. 
It  is  possible  to  raise  the  body  temperature  as  much  as  3  or  4  deg  C  by  this  means.  The 
method  has  proved  useful  in  the  treatment  of  syphilis,  pleurisy,  neuritis,  and  other  diseases. 
The  currents  normally  used  range  from  0.5  to  5.0  amp. 

6.  HIGH-FREQUENCY  SURGERY 

If  one  large  electrode  is  applied  to  a  part  of  the  body  and  a  second  electrode  consisting 
of  a  large  platinum  needle  is  brought  in  contact  with  any  other  part  of  the  body,  the  re- 
sultant current  density  is  so  high  at  the  point  of  application  of  the  needle  that  the  tissue 
may  be  completely  destroyed  by  heat.  This  method,  which  is  called  electrosurgery,  is  a 
relatively  recent  development.  The  small  blood  vessels  severed  by  means  of  the  high- 
frequency  knife  are  normally  sealed  by  the  heating  action,  and  bleeding  is  therefore  con- 
siderably reduced.  Basically,  there  is  no  difference  between  the  equipment  used  for 
diathermy  and  for  high-frequency  surgery. 

BIBLIOGRAPHY 

Cumber-batch,  E.  P.,  Diathermy.  London  (1929). 

Kova^s,  Richard,  M.  DM  Electro-Therapy  and  Light  Therapy,  Lea  and  Febiger  (1932). 
Osbourne,  S.  L.,  and  H.  J.  Holmquest,  Technic  of  Electro-Therapy,  and  Its  Physical  and  Physiological 
Bases.     Charles  C.  Thomas,  Springfield,  111.,  pp.  315-744  (1944). 

THE  MEDICAL  USES  OF  ULTRAVIOLET 
AND  INFRARED  RADIATIONS 

7.  APPARATUS 

Apparatus  for  the  production  of  ultraviolet  radiation  includes  the  carbon-arc  generator, 
the  quartz-enclosed  mercury-arc  generator,  and  the  mercury  glow-discharge  lamp  enclosed 
in  quartz  or  Corex,  a  glass  which  transmits  a  useful  amount  of  ultraviolet  radiation.  One 
jof  the  chief  objections  to  the  carbon-arc  generator  is  its  production  of  relatively  high- 
intensity  visible  and  infrared  radiations.  In  addition  to  the  above-mentioned  lamps, 
.which  have  relatively  high  outputs,  several  special  lamps  have  been  developed  for  home 
iase  under  medical  supervision. 


THERAPEUTIC  USE  OF  ULTRAVIOLET  RADIATION      23-07 

One  of  these — the  Mazda  US  lamp — is  rated  at  110  volts  275  watts.  At  a  distance  of 
24  in.  this  lamp  will  produce  a  mild  reddening  (erythema)  of  the  skin  in  about  5  min.  The 
lamp  is  strong  in  those  radiations  that  produce  sunburn  (2800  to  3200  angstrom  units). 
There  is  also  a  relatively  strong  infrared  beam  produced  chiefly  by  a  tungsten  filament  in 
an  atmosphere  of  argon  and  nitrogen. 

The  outer  envelope  of  the  Mazda  RS  lamp  is  made  of  special  glass  that  transmits  in- 
frared and  visible  radiation,  and  ultraviolet  radiation  of  wavelengths  greater  than  2800  A. 
A  reflecting  surface  is  deposited  on  the  interior  surface  of  this  envelope;  the  radiation  is 
emitted  through  the  circular  (12-cm  diameter)  end  of  the  lamp  opposite  the  screwplug. 
A  tungsten  filament,  which  operates  as  a  series  ballast  after  the  initial  preheating  period, 
is  mounted  in  the  outer  envelope;  the  space  is  filled  with  argon  and  nitrogen. 

Within  the  larger  outer  envelope,  there  is  a  quartz  capsule  containing  an  oxide-coated 
filament  and  a  second,  initially  cold,  electrode;  this  capsule  contains  mercury. 

When  the  lamp  is  initially  connected  to  110-volt  60-cycle  alternating  current,  the  outer 
(tungsten)  filament  and  the  inner  (oxide-coated)  filament  are  connected  in  series.  After 
approximately  30  sec,  when  the  oxide-coated  filament  acquires  a  sufficiently  high  tem- 
perature for  copious  emission  of  electrons,  a  thermal  delay  switch  within  the  tube  operates 
to  connect  the  initially  cold  electrode  of  the  quartz  capsule  to  the  terminal  of  the  supply 
opposite  from  the  terminal  connected,  through  the  tungsten  filament,  to  the  oxide-coated 
filament  and  disconnects  the  oxide-coated  filament.  The  mercury  arc  then  strikes,  be- 
coming stable  as  the  initially  cold  electrode  becomes  heated  by  bombardment.  The  arc 
in  the  mercury  vapor  at  a  pressure  of  approximately  1.1  atmospheres  then  operates  in 
series  with  the  tungsten  filament;  the  filament  and  arc  emit  ultraviolet,  visible,  and  in- 
frared radiation.  The  manufacturer  recommends  a  1-min.  "warm-up"  period  before  use. 

Glow-discharge  mercury  tubes  operating  at  voltages  as  high  as  5000  volts  and  currents 
of  the  order  of  15  ma  produce  radiation  almost  entirely  of  the  emission  frequency  line  of 
mercury  having  a  wavelength  of  2537  A.  This  radiation  is  strongly  bactericidal.  Since 
it  also  requires  low-power  input,  this  type  of  lamp  is  often  used  to  irradiate  the  area  in 
operating  rooms  and  in  other  places  where  air-borne  bacteria  are  to  be  mimmized.  Ex- 
periments using  these  devices  in  schoolrooms  are  being  conducted. 

Since  the  mercury-arc  lamp  enclosed  in  quartz  produces  a  relatively  high-intensity  radi- 
ation at  wavelengths  from  2483  to  4047  A,  and  since  this  apparatus  is  relatively  stable 
in  operation  after  a  few  minutes  of  operation,  it  is  now  the  most  common  source  for  med- 
ical purposes.  The  total  power  input  to  such  a  device  is  from  250  to  400  watts,  and  its 
output  of  ultraviolet  radiation  is  of  the  order  of  225  microwatts  per  square  centimeter  at  a 
distance  of  1  meter  without  a  reflector. 

Special  photoelectric  cells  connected  to  integrating  circuits  and  counters  have  been 
devised  to  measure  the  outputs  of  ultraviolet  generators  throughout  various  bands  of 
wavelengths.  However,  such  standardizing  measurements  have  not  been  very  often 
applied  to  clinical  use. 

Infrared  radiation  is  produced  chiefly  by  electrically  heated  wire  operating  at  temper- 
atures from  800  to  1600  deg  C  or  by  means  of  incandescent  tungsten  lamps  with  or  without 
filters. 

8.  THERAPEUTIC  USE  OF  ULTRAVIOLET  RADIATION 

It  has  been  known  for  many  years  that  sunlight  has  a  definitely  beneficial  effect  upon  the 
human  body.  {Since  1800  the  spectra  of  light  emitted  by  the  sun  and  by  electric  arcs  have 
been  analyzed  and  more  attention  has  been  directed  to  the  specific  photochemical  effects 
of  the  various  parts  of  the  spectrum.  More  or  less  arbitrarily,  ultraviolet  radiation  has 
been  defined  as  radiation  with  a  range  of  wavelengths  from  approximately  4000  to  about 
40  A  (1  angstrom  unit  =  10~8  crn).  The  solar  spectrum  extends  down  to  approximately 
2900  A  but  under  average  atmospheric  conditions  at  sea  level  there  is  negligible  radiation 
below  3000  A.  It  has  been  shown  that  most  of  the  ultraviolet  radiation  incident  upon  the 
human  body  from  the  sun  or  from  artificial  sources  is  absorbed  in  the  surface  tissues.  The 
penetration  is  0.1  mm  or  less. 

There  has  been  much  speculation  as  to  the  biological  processes  produced  by  ultraviolet 
radiation,  which  has  so  far  resulted  largely  in  contradictions  and  disagreements  among 
authorities  on  the  subject. 

The  first  noticeable  effects  of  ultraviolet  radiation  are  erythema,  or  reddening  of  the 
skin,  usually  followed  by  pigmentation.  Each  individual  has  a  different  tolerance  to 
ultraviolet  radiation.  What  may  cause  a  pronounced  erythema  in  one  may  produce  no 
effect  upon  another.  It  has  been  discovered  that  certain  non-soluble  fatsr  particularly 
ergosterol,  form  vitamin  D  when  irradiated  by  ultraviolet  rays.  This  substance  is  present 
in  the  human  body,  and  normal  exposure  to  sunlight  is  one  method  for  supplying  the 


23-08  MEDICAL  APPLICATIONS  OF  ELECTRICITY 

definite  need  of  the  body  for  vitamin  D.  Although  there  is  no  question  as  to  the  beneficial 
effects  of  sunlight  and  artificial  sources  of  ultraviolet  radiation  when  exposures  are  care- 
fully controlled,  enthusiasm  on  the  part  both  of  physicians  and  of  laymen  has  led  to  dan- 
gerous overexposures  to  both  natural  and  artificial  sources  of  this  form  of  energy.  It  has 
not  been  proved  that  exposure  to  direct  sunlight  is  essential  to  the  physical  well-being  of 
the  normal  healthy  human. 

Treatment  by  ultraviolet  radiation  is  useful  in  cases  of  rickets,  high  blood  pressure,  and 
some  skin  diseases.  It  has  been  shown  definitely  that  ultraviolet  radiation  produces 
directly  and  indirectly  a  substantial  rise  in  the  amount  of  calcium  and  phosphorus  in  the 
blood.  The  treatment  is  in  general  use  for  rickets  in  infants  and  has  been  remarkably 
successful.  Certain  experimenters  believe  that  the  effect  of  ultraviolet  rays  on  the  nervous 
system  is  stimulating.  A  number  of  psychiatrists  have  recommended  the  treatment  for 
neurasthenia  and  some  of  the  psychoses. 

£  After  Pasteur's  discovery  that  small  living  organisms  are  the  cause  of  many  diseases 
and  also  of  fermentation,  experiments  were  performed  to  find  the  effects  of  various  types 
of  radiation  upon  these  organisms.  It  was  found  that  ultraviolet  rays  were  potent  in 
bactericidal  action,  particularly  in  the  2600  A  region. 

Different  parts  of  the  body  vary  considerably  in  sensitivity.  For  example,  twenty  to 
thirty  times  as  much  ultraviolet  radiation  is  required  to  produce  an  erythema  of  the  soles 
of  the  feet  than  of  the  face,  which  is  the  most  sensitive  part.  Even  small  doses  which 
penetrate  to  the  eyeball  can  cause  serious  damage;  therefore  all  patients  receiving  ultra- 
violet treatment  are  required  to  wear  goggles  made  of  glass  which  absorbs  practically  all 
the  ultraviolet  radiation.  Practically  no  ultraviolet  radiation  is  transmitted  through 
window  glass  and  very  little  through  smoky  atmosphere.  Therefore  sunlight  may  be  used 
as  a  source  of  therapeutic  ultraviolet  rays  to  the  best  advantage  in  special  regions — for 
example,  in  high  mountains  where  there  are  generally  very  few  clouds  and  practically  no 
dust  or  smoke.  Since  the  ultraviolet  content  of  sunlight  varies  with  latitude,  season,  time 
of  day,  and  atmospheric  conditions  (such  as  ozone,  water  vapor,  dust,  and  smoke)  it  is  im- 
portant to  measure  the  intensity  in  the  region  of  2900-3200  A  during  treatment. 

9.  THERAPEUTIC  USE  OF  INFRARED  RADIATION 

Infrared  radiation  is  used  for  surface  heating,  either  of  a  small  circumscribed  area  or  for 
the  surface  of  an  entire  leg  or  arm.  One  of  the  chief  effects  is  the  production  of  increased 
blood  flow  near  the  surface,  and  this  may,  in  turn,  cause  more  deep-seated  changes. 

It  is  possible  also  to  change  the  temperature  of  the  entire  body.  Thus  the  body  may 
be  placed  in  a  heat-insulated  housing  inside  of  which  incandescent  lamps  or  other  heating 
elements  raise  the  temperature  of  the  body  and  of  the  area  surrounding  it.  Since  there  is 
no  way  for  this  heat  to  be  entirely  radiated,  the  temperature  of  the  body  is  caused  to  rise. 
Such  whole-body  heating  may  also  be  accomplished  by  means  of  diathermy. 

BIBLIOGRAPHY 

Bernhard,  O.,  Light  Treatment  in  Surgery.     Arnold,  London  (1923). 

Duggar,  B.  A.,  Biological  Effects  of  Radiation,  Vol.  1.     McGraw-Hill  (1936). 

Finsen,  N.  R.,  Phototherapy.     Arnold,  London  (1901). 

Glasser,  Otto,  Editor,  Medical  Physics.     Year  Book  Publications,  Inc.,  Chicago,  111.  (1944).     Physical 

therapy — heat  and  cold,  pp.  1043-1054;  light,  pp.  1054-1068;  radiation,  sources  of  ultraviolet  and 

infrared,  pp.  1157-1163. 

Laurens,  Henry,  The  Physiological  Effects  of  Radiant  Energy.     Chemical  Catalog  Co.  (1923). 
Bollier,  A.,  Heliotherapy.     Frowde,  Hodder  and  Stoughton,  London  (1923). 
Osbourne,  S.  L.,  and  H.  J.  Holmquest,  Technic  of  Electro-therapy  and  Its  Physical  and  Physiological 

Bases.     Charles  C.  Thomas,  Springfield,  111.  (1944).     Ultraviolet  radiation,  pp.  235-313;  infrared 

radiation,  pp.  194-234. 


ELECTROCARDIOGRAPHY  AND  ELECTRO- 
ENCEPHALOGRAPHY 

10.  APPARATUS 

Muscular  contractions  and  the  functioning  of  nerves  are  accompanied  by  measurable 
differences  of  electric  potential  at  the  surface  of  the  body.  Thus  the  time-varying  dif- 
ferences of  potential  between  the  right  arm  and  the  left  arm,  the  right  arm  and  the  left  leg, 


ELECTROCARJDIOGKAPHY 


23-09 


the  left  arm  and  the  left  leg  are  used  by  cardiologists  to  interpret  the  action  of  the  heart; 
these  differences  of  potential  are  called  respectively  the  potentials  of  Lead  I,  Lead  II,  and 
Lead  III.  The  measured  differences  of  potential  are  of  the  order  of  a  few  millivolts.  An 
electrocardiogram  for  Lead  II  for  a  normal  patient  and  a  heart  sound  record  made  simul- 
taneously are  shown  in  Fig.  1.  For  the  electrocardiogram,  a  vertical  deflection  of  one 
small  division  corresponds  to  a  difference  of  potential  of  0.2  mv. 


I® 

fa  i-      ~^~ 

V  /?N 

V  I 

©            ® 

J 

©    ©       i 

FIG.  1.    Electrocardiogram 

Lower. — Electrocardiogram,  Lead  II,  normal  patient.  The  letters  P,  Q,  R,  S,  and  T  are  standard 
symbols  for  designating  the  waves  thus  marked. 

Upper. — Heart  sound  record  made  simultaneously  with  the  electrocardiogram  above. 

Electrocardiographic  apparatus  requires  the  following  component  parts:  (1)  a  device 
for  transforming  the  electromotive  force  generated  by  the  heart  into  the  motion  of  a  light 
beam  or  of  a  shadow;  (2)  a  camera  with  moving  film  or  paper  to  record  the  deflections  of 
the  light  beam  or  shadow;  (3)  devices  for  calibrating  the  abscissa  (time)  and  the  ordinate 
(electromotive  force)  on  the  electrocardiogram. 

Einthoven  developed  a  system  similar  to  that  shown  in  Fig.  2.  A  silvered  quartz  string 
with  variable  tension  is  held  perpendicular  to  an  intense  constant  magnetic  field  produced 
by  an  electromagnet.  A  calibrated  ex- 
ternal source  of  electromotive  force  is 
used  as  standardizes  A  variable  resist- 
ance and  associated  battery  are  used  to 
compensate  for  skin  currents  (described 
below  under  technique).  A  specially 
adapted  camera,  with  a  tuning  fork  or 
other  timing  device,  is  used  to  record 
the  deflection  of  the  string  produced  by 
the  heart  action. 

It  has  been  shown  that  the  Einthoven 
string  galvanometer  produces  distortion 
due  to  the  change  in  its  response  with 
frequency;  furthermore,  as  is  described 
below,  it  requires,  for  an  adjustment  of 

sensitivity,  a  change  in  the  string  ten-  2     Outline  Diagram  of  Einthoven  String  Electro- 

sion  which  produces  further  distortion.  cardiograph 

Therefore,  efforts  were  made  to  use  lf  galvanometer  conducting  string  of  silvered  quartz, 
galvanometers  with  more  uniform  fre-  suspended  with  controllable  tension  between  the  poles 
quency  response.  It  was  found  that  of  an  electromagnet  (not  shown)  which  produces .a  mag- 
j,  ,  AH  .  ,  j  ,T_  netic  field  in  the  direction  of  the  arrow  2;  3,  light  source 

the  use  of  carefully  constructed  ther-  to  Obtam,  with  system  of  lenses  not  shown,  an  image  of 
mionic  amplifiers  in  association  with  the  string  upon  the  moving  photographic  film  in  the 
these  new  galvanometers  provided  in-  ^^'i^^^^M*  S£S±£S! 
struments  of  great  flexibility  ana  sta-  a  series  circuit  which  introduces  an  adjustable  electro- 
bility.  The  cardiographic  p'aper  or  film  motive  force  in  series^  with  the  string  to  compensate  for 
can  be  driven  by  a  synchronous  motor, 
so  that  the  time  abscissas  are  the  same 
for  all  cardiograms. 


leads  to  the  patient. 


It  is  most  convenient  to  describe  the  technique  of  electro  car  diography  by  referenee^to 
the  Einthoven  instrument  shown  in  Fig.  2.  A  resistance  5  is  connected  in  parallel  with 
the  string,  reducing  its  sensitivity  by  a  ratio  of  10  to  1.  One  area  on  each  arm  of  the 
patient  is  rubbed  with  a  contact  paste  and  a  metal  electrode  is  applied.  This  corresponds 
to  the  designation  described  above  as  Lead  I.  A  switch  is  closed  connecting  the  patient 
to  the  electrocardiograph.  The  galvanometer  deflects,  owing  to  what  is  called  skin  cur- 
rent. The  slide  of  9  is  then  moved  until  the  galvanometer  string  is  returned  to  its  zero 
position.  This  operation  balances  out  the  skin  current,  which  is  constant  and  plays  no 
part  in  the  interpretation  of  the  electrocardiogram.  After  the  compensator  is  adjusted, 


23-10 


MEDICAL  APPLICATIONS  OF  ELECTRICITY 


the  shunt  5  across  the  galvanometer  string  is  disconnected.  This  increases  the  amplitude 
of  swing  of  the  fiber.  The  standardizing  circuit  is  then  connected  by  closing  switch  14 
and  opening  switch  11.  The  circuit  12,  13,  15  is  designed  to  produce  a  potential  drop  of 
1  mv  across  the  resistance  13.  This  throws  the  image  of  the  string  across  the  screen.  An 
adjustment  of  the  mechanical  tension  of  the  string  is  made  so  that  the  application  of  the 
1-mv  standardizing  voltage  produces  a  deflection  of  exactly  1  cm.  The  standardizing 
voltage  is  then  removed  by  opening  switch  14  and  closing  switch  11,  and  the  electro- 
cardiogram for  Lead  I  is  taken  by  means  of  a  paper  moving  at  constant  speed  past  the 
beam  of  light  through  the  galvanometer.  For  each  centimeter  deflection  of  this  record 
an  electrocardiographic  impulse  of  1  mv  is  necessary.  The  same  procedure  is  followed  for 
Leads  II  and  III,  The  moving  photographic  paper  passes  before  a  glass  screen  upon 
which  lines  1  mm  apart  are  ruled  in  the  direction  of  motion  of  the  paper,  while  a 
synchronous  motor  turns  a  bladed  wheel  through  the  light  beam  at  right  angles  to  the 
direction  of  motion  of  the  paper,  producing  a  series  of  time-marking  lines  spaced  at 
0.04-sec  intervals. 

The  procedure  described  can  be  accomplished  quickly  in  practice.  The  entire  operation 
is  illustrated  in  Fig.  3.  This  description  applies  to  the  older  type  of  electrocardiograph, 

from  which  the  underlying  principles  can  be  clearly 
understood.  The  routine  use  of  a  modern  electro- 
cardiograph is  characterized  by  practical  simplic- 
ity, although  a  description  of  this  procedure  would 
not  illustrate  so  clearly  the  fundamental  charac- 
teristics. 

The  modern  electrocardiograph  may  be  battery 
operated  or  it  may  operate  from  a  110-volt  60-cycle 
source.  It  is  usually  a  self-contained  portable  de- 
vice, and  the  records  may  be  produced  on  film  or 
photographic  paper  or — more  recently — may  be 
directly  recorded  on  paper  that  may  be  examined 
as  soon  as  it  is  produced.  These  devices  are 
equipped  with  stabilized  audio-frequency  ampli- 
fiers feeding  into  specially  designed  galvanometers. 
The  frequency  characteristic  of  these  electro- 
cardiographs results  in  the  reproduction,  at  nearly 
constant  levels,  of  frequencies  from  0.5  to  50  or 
100  cycles  per  second.  Because  of  the  susceptibility  of  these  high-impedance  amplifiers 
to  hum  pickup,  special  hum-bucking  input  circuits  are  incorporated. 

An  electroencephalograph  is  used  to  record  time-varying  changes  of  potential  between 
pairs  of  electrodes  in  contact  with  various  parts  of  the  scalp;  the  magnitudes  of  these  dif- 
ferences of  potential  are  from  a  few  microvolts  to  approximately  100  mv.  The  component 
frequencies  of  the  electroencephalographic  signals  are  less  than  1000  cycles  per  second. 
It  follows,  therefore,  that  an  electroencephalograph  can  be  a  device  similar  to  an  electro- 
cardiograph except  that  (1)  the  amplification  must  be  about  100  times  greater  in  the 
electroencephalograph,  and  (2)  it  is  considered  essential  to  provide  fourteen-channel  in- 
put (and  two-  to  six-channel  output)  for  the  electroencephalograph  instead  of  four-channel 
input  which  is  used  in  the  electrocardiograph. 


.!3    lliUClbl  OibCU.    J.JUL    JL'  J.g.     O.  JL.JJL.Li3     U.O»l^J.AjJI 

•ill 


FIG.  3.    Calibration  of  Electrocardiograph 

The  calibration  record  which  is  made  at 
the  t  end  of  the  electrocardiogram  of  each 
patient.  At  (1)  the  standardizing  voltage 
of  1  mv  is  connected  in  series  with  the  gal- 
vanometer, causing  a  deflection  of  1  cm  to 
(2).  The  electrocardiogram  is  of  Lead  II, 
normal  patient. 


11.  TECHNIQUES 

Since  the  middle  of  the  nineteenth  century  it  has  been  known  that  an  electromotive 
force  is  generated  within  the  heart  during  the  period  of  contraction  of  this  muscular  organ. 
The  precise  causes  of  this  effect  are  unknown.  Nevertheless,  the  methods  for  measuring 
the  variations  of  this  electromotive  force  with  time  have  been  carefully  standardized,  and 
statistical  records  have  been  made  for  many  years.  By  associating  these  records  with 
case  histories,  a  technique  has  been  evolved  for  diagnosing  certain  diseases  of  the  heart 
which  is,  in  many  instances,  remarkably  valuable.  The  development  of  electrocardiog- 
raphy  has  shown  the  benefits  of  carefully  standardized  scientific  methods  more  than  most 
other  special  fields  of  medicine  in  which  electrical  equipment  has  been  used. 

When  the  so-called  "brain  waves"  are  recorded  by  means  of  an  electroencephalograph 
the  wave  forms  are  found  to  vary  with  age,  with  the  somatic  state  of  the  individual,  with 
the  state  of  mental  health  of  the  individual,  and  with  many  other  factors.  Thus  the  de- 
velopment of  this  field  of  investigation  has  been  based  upon  the  recording  of  many  elec- 
.trpencephalograms,  their  analysis  and  intercomparison,  and  finally  attempts  at  diagnosis 
^ased,  >  upon  these  analyses.  The  electroencephalograph  now  finds  clinical  use  in  the 
corroborative  diagnosis  of  epilepsy  and,  in  many  cases,  in  the  localization  of  brain  tumors. 


THE   STETHOPHONE  23-11 

BIBLIOGRAPHY 

A8fe^'  Richard-  and  Edgar  Hull,  Essentials  of  Electrocardiograph,  The  MacmiUan  Co.,  New  York 

(.1945;. 

Berger,  EC.,  Archives  of  Psychiatry,  Vol.  87,  527  (1929);  Vol.  100,  301  (1933). 
Glasser,  Otto,  Editor,   Medical  Physics.     The  Year  Book  Publications,  Inc.,  Chicago,  111.   (1944). 

Electrocardiography,  pp.  352-360;  electroencephalography,  pp.  361-371. 
Jasper,  H.  H.,  and  L.  Carmichael,  Science,  Vol.  81  (January  1935). 
Lilcotf,  W    J3     M.  B.  Rappapart,  and  S.  A.  Levine,  Continuous  Recording  Electrocardiography,  Am. 

Heart  /.,  Vol.  28,  98-114  (July  1944). 
Traugott,  Paul,  Electroencephalograph  Design,  Electronics,  Vol.  16,  132-144  (August  1943). 

ELECTROACOUSTIC  DEVICES 

12.  AIDS  TO  THE  DEAF 

Those  whose  hearing  is  deficient  require  aids  designed  upon  the  basis  of  a  quantitative 
study  of  their  relative  deafness.  For  this  study  a  device  called  the  audiometer  is  used; 
this  instrument  consists  essentially  of  a  vacuum-tube  oscillator  capable  of  producing 
practically  pure  tones  accurately  controllable  as  to  pitch  and  intensity  in  head  receivers. 
The  pitch  proceeds  by  octaves  from  32  double  vibrations  per  second  to  16,384,  with  an 
intensity^range  of  109.  Charts  called  audiograms  have  been  made,  showing  the  thresholds 
in  sensation  units,  of  hearing  and  of  pain,  for  thousands  of  subjects  with  normal  hearing. 
By  comparing  the  audiograms,  both  the  type  and  degree  of  deafness  may  be  estimated. 
This  constitutes  a  valuable  aid  in  diagnosis  and  also  is  a  useful  adjunct  for  the  prescription 
of  hearing  aids. 

Many  who  are  deaf  learn  to  read  lips.  Care  must  be  taken,  therefore,  to  eliminate  this 
source  of  error  from  tests  of  hearing  aids,  even  though  lip-reading  may  become  a  part  of 
the  method  of  understanding  speech  after  the  hearing  aid  is  adopted.  Some  who  have 
beon  deaf  for  a  period  of  years  require  a  considerable  amount  of  time  to  learn  to  under- 
stand sounds  heard  with  the  aid  of  an  electric  device. 

Hearing  aids  usually  consist  of  a  small  microphone,  a  battery,  and  a  special  light-weight 
electromagnetic  or  crystal  receiver.  The  system  is  designed  to  produce  at  the  ear,  as 
nearly  as  possible,  an  amplified  replica  of  the  sounds  incident  upon  the  microphone.  If 
such  a  simple  device  does  not  amplify  sufficiently,  an  amplifier  using  vacuum  tubes  may 
be  added. 

In  recent  years,  the  development  of  so-called  "miniature"  and  "hearing-aid"  tubes  and 
of  small  batteries  and  printed  circuits  has  considerably  reduced  the  size  and  weight  of 
such  equipment. 

More  elaborate  devices  have  been  made  for  those  whose  hearing  is  especially  deficient, 
in  some  instances  the  apparatus  being  designed  to  suit  the  individual  requirements  as 
determined  from  studies  of  the  audiograms  and  the  personal  characteristics  of  the  subject. 
Among  the  more  elaborate  examples  of  this  type  of  hearing  aid  is  the  two-channel  system 
in  which, two  high-fidelity,  velocity  microphones  are  connected  individually  to  two  high- 
quality  amplifiers,  equipped  with  filters  especially  designed  to  compensate  for  the  auditory 
deficiencies  of  the  subject.  The  outputs  of  these  two  systems  are  then  connected  to  in- 
dividual headphones  so  that  each  ear  of  the  subject  has  an  entirely  separate  channel, 
separately  connected,  and  affording  corrected  binaural  (two-ear)  hearing. 

13.  THE  STETHOPHONE 

About  1924  a  device  for  picking  up  human  heart  sounds  by  means  of  a  microphone  and 
amplifying  these  sounds  was  developed  and  given  the  name  "stethophone."  The  original 
purpose  of  the  device  was  to  make  it  possible  for  students  in  the  amphitheater  of  the  heart 
clinic  to  listen  simultaneously  with  the  demonstrator  to  the  heart  sounds  of  the  patient 
under  examination.  The  device  had  the  additional  advantage  over  the  ordinary  stetho- 
scope of  producing  louder  sounds  in  the  listener's  ear.  It  was  further  shown  that  by  the 
insertion  of  electrical  filters  it  was  possible  to  eliminate  portions  of  the  audible  frequency 
spectrum  in  order  to  concentrate  upon  certain  specific  sounds.  This  means  of  selective 
listening  proved  an  aid  in  the  diagnosis  of  murmers  and  other  cardiac  abnormalities. 
More  recently  this  device  has  been  used  by  physicians  who  are  somewhat  hard  of  hearing. 

RECORDING  OF  HEART  SOUNDS.  Heart  specialists  have  also  shown  that  it  is  of 
value  to  record  the  heart  sounds  on  a  moving  strip  of  paper  or  film  simultaneously  with  an 
electro  car  diographic  record.  This  dual  record  has  proved  an  aid  in  the  diagnosis  of  certain 
cardiac  disorders,  and  also  for  purposes  of  research  into  the  mechanism  of  the  heart. 


23-12  MEDICAL  APPLICATIONS  OF  ELECTRICITY 


BIBLIOGRAPHY 

Cabot,  R.  C.,  and  H.  F.  Dodge,  Frequency  Characteristics  of  Heart  and  Lung  Sounds,  /.  Am.  Medical 

Assoc.,  Vol.  84,  24  (June  1945). 

Fletcher,  Harvey,  Speech  and  Hearing.     Van  Nostrand  (1929) . 

Frederick,  H.  A.,  and  H.  F.  Dodge,  The  Stethophone,  Bell  Sys.  Tech.  J.,  Vol.  3,  531-549  (October  1924). 
Gamble,  C.  J.,  and  D.  E.  Replogle,  A  Multiple  Electrical  Stethoscope  for  Testing,  J.  Am.  Medical  Assoc., 

Vol.  82,  388  (February  1924). 

Margolis,  A.,  Archives  of  Internal  Medicine,  Vol.  46,  1048  (December  1930). 
Rappapart,  M.  B.,  and  H.  B.  Sprague,  Graphic  Registration  of  Heart  Sounds,  Am.  Heart  J.,  Vol.  23, 

591-623  (May  1942). 
Williams,  H.  B.,  and  H.  F.  Dodge,  Analysis  of  Heart  Sounds,  Archives  of  Internal  Medicine,  Vol.  38, 

685-693  (December  1926). 

ROENTGEN  THERAPY 

14.  PURPOSE  AND  GENERAL  TECHNICAL  REQUIREMENTS 

A  short  time  after  the  announcement  by  Roentgen  of  the  discovery  of  x-rays  in  1895, 
a  number  of  physicians  began  to  use  the  new  radiation  for  the  treatment  of  certain  diseases. 
It  has  been  only  since  1920,  however,  that  important  detailed  data  have  been  recorded  and 
correlated.  It  has  been  definitely  determined  that  the  absorption  of  x-rays  by  living 
tissue  can  cause  the  destruction  of  that  tissue.  Destruction  of  a  given  type  of  tissue  is 
dependent  upon  the  dose  rate  (in  roentgens  per  minute;  see  below),  the  duration  and 
frequency  of  treatments,  the  area  of  the  irradiated  skin  surface  (portal),  the  total  dose, 
and  probably  other  factors  as  well.  The  destructive  action  is  radically  different  with  dif- 
ferent kinds  of  tissue.  The  reproductive  cells  of  the  human  body  are  most  sensitive,  and 
the  bones  are  least  affected.  The  treatment  of  malignant  tumors,  such  as  cancer,  is  based 
upon  the  fact  that,  to  a  certain  extent,  it  is  possible  to  cause  destruction  of  the  diseased 
tissue  without  permanently  damaging  normal  surrounding  tissue. 

X-rays  generated  by  means  of  a  hot-cathode  tube  have  continuous  spectra  character- 
ized by  a  minimum  wavelength  dependent  upon  the  maximum  x-ray  tube  voltage.  The 
shorter  the  wavelength,  the  more  penetrating  is  the  radiation  (see  also  Section  4,  x-ray 
tubes).  For  these  reasons  the  kind  of  x-ray  equipment  chosen  for  therapeutic  use  de- 
pends upon  the  site  of  the  diseased  portion  which  it  is  intended  to  treat.  If,  for  example, 
it  is  desired  to  treat  the  skin,  it  may  prove  desirable  to  use  radiation  of  a  wavelength  of 
the  order  of  1  A.  Such  radiations  are  sometimes  called  Grenz  rays.  The  most  generally 
used  x-ray  therapy  apparatus  is  operated  at  x-ray  tube  voltages  of  50  to  400  kvp  (peak 
kilo  volts).  A  filter  consisting  of  a  few  millimeters  of  aluminum,  a  few  tenths  of  a  milli- 
meter of  copper,  or  a  combination  of  the  two,  is  inserted  between  the  x-ray  tube  and  the 
patient.  This  filter  absorbs  a  large  part  of  the  low-frequency  radiation  which  would  other- 
wise be  absorbed  by  the  skin.  The  filtered  x-ray  energy  penetrates  to  the  site  of  disease, 
and  a  reasonable  proportion  is  absorbed  and  helps  to  produce  the  desired  effect.  Treat- 
ments by  x-ray  therapy  have  been  apparently  beneficial  in  some  cases,  particularly  if  the 
diseased  part  is  properly  diagnosed  early  in  its  development. 

The  gamma  rays  of  radium  have  been  valuable  in  treating  cancer.  It  has  therefore  been 
assumed  that  x-ray  apparatus  capable  of  generating  radiation  comparable  in  frequency 
to  gamma  rays  might  prove  useful.  Experiments  in  this  direction  have  led  to  the  con- 
struction of  tubes  and  apparatus  capable  of  operating  at  600  kvp  to  2  Mev. 

APPARATUS.  The  high-voltage  generator  (50-400  kvp)  required  for  exciting  an 
x-ray  therapy  tube  is  usually  constructed  with  one  of  the  following  typical  circuits  as  a 
basis  (see  also  Section  7,  Power  Supply) :  (1)  half-wave  thermionic  rectification;  (2)  special 
half-wave  rectified  voltage-doubling  circuits  with  condensers  (see  Fig.  1) ;  (3)  full-wave 
thermionic  rectification  with  condensers  (nearly  constant  potential) . 

Two  kinds  of  apparatus  have  been  devised  and  put  into  relatively  common  use  for  oper- 
ation at  1  and  2  Mev. 

In  one  of  these  devices  a  van  de  Graaff  generator  and  an  x-ray  tube  are  assembled  in  a 
steel  tank  into  which  air  is  introduced  at  a  pressure  of  several  atmospheres,  the  spark- 
over  potential  gradient  of  gases  at  high  pressures  being  considerably  greater  than  at  nor- 
mal atmospheric  pressure.  The  van  de  Graaff  generator  consists  of  a  continuous  belt  of 
insulating  material  mounted  on  two  pulleys,  one  of  which  is  at  ground  potential  and  is 
driven  by  a  motor.  Near  the  bottom  pulley  electrodes  are  mounted  in  front  of  and  in 
back  of  the  belt.  A  difference  of  potential  is  applied  to  these  electrodes  so  that  negative 
charges  (electrons)  are  deposited  on  the  belt.  Thus  the  moving  belt  effectively  carries  a 
negative  charge  toward  the  top  pulley.  Surrounding  this  upper  pulley  there  is  mounted 


ROENTGEN   THERAPY 


23-13 


a  hollow  metal  electrode  to  which  a  brush  near  the  belt  is  connected.  Electrons  from  the 
belt  are  conducted  to  the  exterior  surface  of  the  hollow  electrode,  which  thus  acquires  a 
negative  charge.  The  cathode  of  the  x-ray  tube  is  connected  to  this  upper  electrode  of 
the  generator;  the  anode  of  the  tube  is  grounded.  In  order  to  maintain  uniform  potential 
gradient  throughout  the  length  of  the  tube,  cylindrical  accelerating  anodes  are  sealed  into 
the  glass  column  of  the  tube  and  connected  to  taps  on  a  resistor  which  in  turn  is  connected 
in  parallel  with  the  generator.  The  potential  of  the  upper  electrode  with  respect  to  ground 
increases  at  the  beginning  of  operation  until  the  sum  of  the  resistor  current,  the  x-ray  tube 
£Uu  T.U  ,u6  !a1kage  currents  is  equal  to  the  rate  at  which  charge  is  carried  up  on  the 
J-nus  the  tube  operates  at  constant  potential  at  voltages  from  1  to  2  Mev,  and 
currents  of  the  order  of  a  few  tenths  milliampere  are  obtained. 

The  second  type  of  equipment  used  for  voltages  above  1  Mev  employs  the  same  kind 
of  x-ray  tube  as  the  device  described  above  but  a  different  kind  of  high-voltage  generator. 
The  generator  consists  of  an  auto  transformer  (air  core)  operating  at  180  cycles  per  second. 


FIG.  1.    X-ray  Therapy  Apparatus,  with  Half-wave,  Voltage-doubling  Circuit 

5.  a-c  leads;  6,  auto-transformer;  8,  a-c  voltmeter;  9,  exposure  timer;  10.  high-tension  transformer; 
13,  thermionic  valve;  14,  d-o  milliammeter;  15,  x-ray  tube  of  which  16  is  the  anode,  17,  the  cathode;  IS, 
cathode  a-c  ammeter;  19,  the  cathode  heating  transformer;  20,  the  cathode  current  regulator;  and 
27,  the  condenser. 

The  primary  consists  of  a  few  turns  of  wire  near  the  anode  end  of  the  tube  wound  coaxially 
with  the  axis  of  the  tube.  The  secondary  consists  of  a  series  of  coils  (each  connected  to 
an  accelerating  anode  in  the  tube)  mounted  one  above  the  other  along  the  tube  so  that  the 
top  coil  is  approximately  at  the  level  of  the  cathode  of  the  tube.  This  apparatus  is  en- 
closed in  a  steel  tank  into  which  "Freon"  gas  is  pumped  at  a  pressure  of  approximately  3 
atmospheres. 

15.  TECHNIQUE 

The  x-ray  tube  mountings  are  arranged  so  that  the  patient  may  recline  in  a  comfortable 
position  during  treatment.  A  lead-lined  cone  protects  the  patient  and  operator  from 
scattered  radiation.  The  radiation  coming  through  the  filter  upon  the  patient  is  thus 
confined  to  an  area  of  10-400  sq  cm  called  a  portal.  Investigations  have  been  made  to 
determine  how  much  radiation  is  absorbed  at  various  depths  within  the  tissue.  Sometimes 
in  order  to  get  the  desired  absorption  within  a  deeply  seated  tumor  it  is  necessary  to  turn 
the  patient  and  to  give  several  exposures  through  different  portals,  the  central  x-ray  beam 
passing,  in  each  case,  through  the  tumor.  This  method  of  cross-firing  prevents  the  ab- 
sorption in  any  particular  skin  area  from  exceeding  a  tolerable  dose.  The  distance  from 
the  tube  to  the  patient  is  generally  50  to  100  cm.  Tubes  operate  at  currents  of  5  to  30 
ma,  and  the  time  of  exposure  may  be  from  5  to  45  min. 

Just  as  with  ultraviolet  treatment,  the  individual  tolerance  must  be  investigated  to 
prevent  x-ray  burn.  In  general  a  large  factor  of  safety  is  allowed  to  prevent  such  a  pos- 
sibility. 

The  unit  of  x-ray  dosage  is  called  the  roentgen.  It  is  denned  in  terms  of  measurement 
by  means  of  an  ionization  chamber.  The  roentgen  is  the  quantity  of  x-  or  gamma-radia- 
tion such  that  the  associated  corpuscular  emission  per  0.001293  gram  of  air  produces  (in 
air)  ions  carrying  1  esu  of  quantity  of  electricity  of  either  sign.  It  is  assumed  and  ap- 
parently justified  by  empirical  results  that  this  measurement  parallels  the  biologic  effect 
of  x-rays.  The  unit  is  an  international  standard.  In  some  laboratories  an  ionization 


23-14  MEDICAL  APPLICATIONS  OF  ELECTRICITY 

chamber  is  connected  at  all  times;  and  in  a  few,  the  chamber  operates  an  electric  counter 
which  integrates  the  total  dose  and  turns  off  the  power  supply  to  the  x-ray  tube  after  the 
desired  exposure.  In  other  laboratories  the  x-ray  machine  is  calibrated  by  means  of  a 
sphere  gap  and  a  milliammeter  which  measure  respectively  the  peak  kilovoltage  supply  to 
the  x-ray  tube  and  the  average  current  through  the  x-ray  tube.  This  calibration  in  turn 
is  referred  to  the  results  of  ionization-chamber  measurements  made  several  times  each 
year. 

BIBLIOGRAPHY 

Charlton.  E.  E.,  W.  F.  Westendorp,  G.  Hotaling,  and  L.  E.  Dempster,  New  Million-volt  X-ray  Outfit, 
J.  Applied  Phys.,  Vol.  10,  6,  374  (1939). 

Duggar,  B.  A.,  Biological  Effects  of  Radiation.     McGraw-Hill  (1936). 

Glasser,  Otto,  Editor,  Science  of  Radiology.     Charles  C.  Thomas,  Springfield,  111.  (1933). 

Glasser,  Otto,  L.  S.  Taylor,  Edith  Quimby,  and  J.  L.  Weatherwax,  The  Physics  of  Radiology.  Hoeber 
(1944). 

Mayneord,  W.  B.,  The  Physics  of  X-ray  Therapy.     Churchill,  London  (1929). 

Robertson,  J.  K.,  Radiology  Physics.     Van  Nostrand  (1941). 

van  de  Graaff,  R.  J.,  and  J.  G.  Trump,  Design  of  a  Million-volt  X-ray  Generator  for  Cancer  Treatment 
and  Research,  J.  Applied  Phys.,  Vol.  8,  9,  602  (1937). 

Weyl,  Charles,  S.  R.  Warren,  Jr.,  and  D.  B.  O'Neill,  Radiologic  Physics,  Charles  C.  Thomas,  Spring- 
field, 111.  (1941). 


ROENTGENOGRAPHY  AND  ROENTGENOSCOPY 

16.  PURPOSE,  GENERAL  TECHNICAL  REQUIREMENTS,  AND 

TECHNIQUE 

In  the  first  group  of  experiments  performed  by  Roentgen  the  following  x-ray  phenomena 
were  observed. 

•  1.  X-rays,  incident  upon  photographically  sensitive  emulsions,  produced  a  latent  image 
similarly  to  visible  light.  Development  of  the  emulsion  produced  a  darkening  of  the  film 
or  plate  throughout  the  area  traversed  by  x-rays. 

2.  X-rays,  incident  upon  barium  platino  cyanide,  produced  a  visible  (fluorescent)  radi- 
ation. 

3.  X-ray  absorption  was  greater  for  a  given  thickness  of  absorbing  material  in  materials 
of  high  density  than  in  materials  of  low  density.     For  a  given  absorbing  material  x-ray 
absorption  was  found  to  be  greater  as  the  thickness  increased. 

These  three  facts  form  the  bases  of  modern  roentgenography. 

The  physically  measurable  characteristics  of  a  roentgenogram  of  a  part  of  the  human 
body  which  are  most  important  for  medical  diagnostic  purposes  are:  (1)  roentgenographic 
density;  (2)  roentgenographic  contrast;  (3)  roentgenographic  sharpness.  Roentgeno- 
graphic density  and  photographic  density  are  identically  defined  as  the  logarithm  to  the 
base  10  of  the  ratio  of  light  incident  upon  a  particular  area  of  film  to  the  intensity  of  light 
transmitted  through  this  area.  Density  is  a  function  of  x-ray  intensity,  therefore  of  x-ray 
tube  voltage,  distance  of  the  plate  from  the  x-ray  tube,  current  through  the  x-ray  tube, 
time  of  exposure,  and  of  several  less  important  factors,  and  also  of  the  type  of  photographic 
material  and  of  the  method  of  development  and  the  fixing  and  drying  of  this  material. 
Density  is,  of  course,  a  function  of  the  physical  characteristics  of  the  object  which  is  roent- 
genographed.  Roentgenographic  density  may  be  measured  with  a  polarization  photom- 
eter or  a  photoelectric  densitometer,  or,  for  rougher  approximation,  densities  may  be  com- 
pared by  eye.  This  last  method  is  most  unsatisfactory.  Roentgenographic  contrast  is 
the  difference  between  the  two  densities  of  two  areas  of  the  roentgenogram  and  is  there- 
fore a  function  of  the  same  variables  as  roentgenographic  density.  If  it  is  desired  to  per- 
ceive the  difference  produced  roentgenographically  by  tissue  of  nearly  similar  x-ray  ab- 
sorption characteristics  in  the  object  roentgenographed,  then  it  is  important  to  control  the 
technique  of  roentgenography  so  as  to  make  this  difference  clearly  visible  to  the  eye. 
Roentgenographic  sharpness  is  the  ability  of  a  particular  roentgenographic  equipment  to 
reproduce  precisely  borderlines  between  contiguous  but  definitely  different  densities.  In- 
formation deduced  from  the  physical  measurements  of  these  characteristics  is  influenced 
greatly  by  physiological  and  psychological  factors  associated  with  the  viewing  of  roent- 
genograms,  which  factors  have  not  as  yet  been  completely  investigated. 

In  order  to  roentgenograph  any  particular  part  of  the  human  body  a  tube  having  a  very 
Small  focal  spot  is  necessary.  It  is  also  advisable  to  remove  the  film  and  object  from  the 
tube  as  far  as  possible  in  order  to  decrease  distortion  due  to  magnification  of  those  parts  o/ 


KOENTGENOGKAPHY  AND   BOENTGENOSCOPY       23-15 


the  object  not  in  contact  with  the  film.  Since  the  focal  spot  of  the  x-ray  tube  is  not  a 
point  there  will  be  some  loss  in  roentgenographic  sharpness  due  to  its  finite  size.  If  the 
object  to  be  roentgeno graphed  may  be  kept  stationary  for  a  moderately  long  period  of 
exposure,  the  focal  spot  may  be  made  correspondingly  small.  This  method  is  employed 
for  roentgenography  of  teeth  and  bones.  If  the  part  to  be  roentgenographed  is  con- 
tinuously in  motion  a  relatively  short  exposure  is  necessary,  in  order  to  arrest  this  motion 
sufficiently.  Therefore,  to  roentgeno  graph  parts  such  as  the  human  chest  the  x-ray  tube 
focal  spot  must  be  made  correspondingly  larger  in  order  to  dissipate  rapidly  the  energy 
necessary  for  short  exposure  time.  For  roentgenography  of  the  chest  and  heart,  exposures 
of  1/3o  to  */5  sec  are  used.  For  roentgenography  of  other  parts  of  the  body  the  exposure 
time  may  be  from  1/2  to  20  sec. 

To  control  the  contrast  in  the  roentgenograms  the  voltage  of  the  x-ray  tube  is  varied. 
The  x-rays  produced  by  high  voltages  are  more  penetrating  than  those  produced  by  lower 
voltages.  The  voltage  of  the  x-ray  tube  is  generally  measured  by  means  of  a  sphere  gap 
and  is  calibrated,  for  the  particular  current  used,  against  the  primary  voltmeter  reading. 
The  x-ray-tube  current  for  exposures  longer  than  3  sec  is  measured  by  a  d-c  d'Arsonval 
milliammeter.  For  very  short  exposures  a  ballistic  milliampere-second  meter  is  used  to 
measure  the  total  quantity  of  electricity  passing  through  the  x-ray  tube.  The  timer  for 
short  exposures  is  arranged  to  make  and  interrupt  the  primary  current  at  zero  points  of 
the  first  and  last  half  cycle  of  the  exposure.  The  target-film  distance  is  varied,  depending 
upon  all  the  other  factors  involved;  it  generally  has  a  value  between  0.5  meter  and  2  meters. 

Secondary  radiation  from  the  heavier  parts  of  the  human  body  emanates  in  all  direc- 
tions, causing  a  general  fogging  effect  over  the  whole  area  of  the  film  and  therefore  reducing 
the  contrast  so  important  for  accurate  diagnosis.  To  minimize  this,  the  Potter-Bucky 
diaphragm,  consisting  of  a  series  of  parallel  lead  strips  perpendicular  to  the  film  and  sep- 
arated from  each  other  by  non-absorbing  strips,  is  made  to  move  over  the  surface  of  the 
film  during  exposure.  The  lead  strips  effectively  absorb  cross-radiation  (secondary  radi- 
ation) and  therefore  make  better  diagnostic  results  possible.  Table  1  gives  approximately 
the  techniques  required  for  making  roentgenograms  of  various  parts  of  the  human  body. 

Table  1.    Technique  for  Roentgenography  and  Roentgeno scopy 


Focal  Spot- 

X-ray  Tube 

X~ray  Tube 

Exposure 

Intensi- 

Purpose 

film  Distance, 

Peak  Volt- 

Current 

Time, 

fying 

meters 

age,  kvp 

Average,  ma 

seconds 

Screens 

General  roentgenoscopy  .  .  .  ,  

0.5-1.0 

60-90 

2-10 

20-60 

Roentgenography 

Hand      

0.7-1.0 

50-60 

50 

1.5 

No 

Elbow  ,  

0.7-1.0 

40-60 

50 

1.5 

No 

Skull  

0.7-1.0 

60-75 

100 

I 

Yes 

Spine  (use  Bucky  diaphragm)  .  .  . 

1.0 

60-90 

200 

1 

Yes 

Colon  (use  Bucky  diaphragm)  .  .  . 

1.0 

60-90 

100 

1 

Yes 

Chest  

1.25-2.0 

45-85 

30-500 

0.4-0.033 

Yes 

17.  APPARATUS 

Roentgeixoscopic  apparatus  comprises  simply  a  high-tension  transformer  with  the 
secondary  connected  directly  to  the  x-ray  tube  (self-rectification)  and  a  control  apparatus 
for  adjusting  the  x-ray-tube  voltage  and  current  to  predetermined  values  from  60  to  90 
kvp  and  from  2  to  10  ma.  Portable  roentgenographic  machines  are  usually  self-rectified. 
In  order  to  avoid  excessive  voltage  on  the  x-ray  tube  during  the  inverse  half  cycles,  roent- 
genographic apparatus  for  use  at  30  to  500  ma  makes  use  of  thermionic  rectifiers.  Up  to 
100  ma,  a  single  thermionic  rectifier  is  connected  in  series  with  the  x-ray  tube,  resulting  in 
half-wave  rectification.  At  higher  x-ray-tube  currents  (the  usual  ratings  are  200  and  500 
ma) ,  four  thermionic  rectifiers  are  connected  in  a  bridge  circuit  to  produce  full-wave  recti- 
fiers. In  addition  there  are  a  few  roentgenographic  machines  using  three-phase  rectifiers; 
there  are  also  roentgenographic  machines  in  which  a  high-voltage  condenser  with  a  ca- 
pacitance of  0.25  to  1.0  juf  is  charged  and  subsequently  discharged  through  the  tube  with 
effective  exposure  times  of  less  than  l/io  sec. 

Several  pieces  of  auxiliary  equipment  are  essential  for  the  production  of  good  roentgen- 
ograms. Generally  the  film  is  contained  in  a  light-tight  cassette  having  a  front  plate  of 
thin  Bakelite  or  thin  aluminum.  Inside  the  cassette  two  intensifying  screens,  one  on 
either  side  of  the  film,  are  arranged  to  maintain  close  contact  with  the  film  when  the  cas- 
sette is  loaded.  The  x-rays  passing  through  the  screens  excite  fluorescence  in  the  calcium 
tungstate  or  zinc  sulfide  contained  therein,  and  this  fluorescent  light  radiation  records 


23-16  MEDICAL  APPLICATIONS  OF  ELECTRICITY 

upon  the  film.  Screens  now  used  produce  about  95  per  cent  of  the  total  roentgenographic 
density,  the  other  5  per  cent  being  caused  by  the  direct  absorption  of  x-rays  in  the  film 
itself. 

Conventional  techniques  require  the  use  of  14-in.  by  17-in.  films  for  a  roentgenogram  of 
a  chest.  If  it  is  desired  to  make  chest  films  of  many  individuals,  the  cost  is  extremely 
high.  This  has  led  to  the  development  of  photofluorographic  equipment  for  making  x-ray 
surveys  of  the  chest  to  discover  early  lesions  of  tuberculosis  and  other  abnormalities.  In 
this  apparatus  the  x-rays  which  have  traversed  the  patient  impinge  upon  a  fluorescent 
screen  mounted  in  a  light-tight  box,  opposite  which  a  photographic  camera  is  focused  upon 
the  screen.  The  camera  records  the  image  from  the  screen  on  35-mm  or  70-mm  roll  film, 
or  on  4-in.  by  5-in.  flat  film.  Although  the  sharpness  of  the  images  of  such  photofluoro- 
grams  is  inferior  to  the  sharpness  of  a  roentgenogram,  the  results  are  believed  by  many 
radiologists  to  be  accurate  enough  for  surveys. 

In  order  to  standardize  photofiuorographic  techniques,  an  automatic  timing  device  has 
been  developed.  Light  from  a  part  of  the  screen  is  focused  upon  an  electron  multiplier 
tube,  the  output  of  which  is  integrated  and  used  to  operate  a  switch  to  discontinue  the 
exposure  at  the  end  of  the  time  required  for  producing  the  proper  density  on  the  film.  This 
automatic  timing  device  is  now  being  tested  for  possible  application  to  standard  roentgen- 
ographic procedures. 

The  techniques  of  medical  roentgenography  have  been  increasingly  used  and  further 
developed  in  the  examination  of  industrial  products.  Thus  the  equipment  operating  at 
2  Mev  is  capable  of  producing  a  satisfactory  record  on  x-ray  film  of  steel  as  thick  as  12  in. 
The  gamma  rays  from  radium  are  also  utilized  to  make  films  of  metal  parts.  These  are 
extremely  valuable  methods  of  inspecting  the  industrial  products  in  order  to  detect  faults 
within  the  product  without  destroying  the  product. 

BIBLIOGRAPHY 

McNeill,  Clyde,  Roentgen  Technique.     Charles  C.  Thomas,  Springfield,  111.  (1939). 
Robertson,  J.  K.,  Radiology  Physics.     Van  Nostrand  (1941). 

St.  John,  A.,  and  H.  R.  Isenburger,  Industrial  Radiology.     2nd  Ed.,  Wiley  (1943). 
Weyl,  C.,  S.  R,  Warren,  Jr.,  and  D.  B.  O'Neill,  Radiologic  Physics.     Charles  C.  Thomas,  Springfield, 
111.  (1941). 

18.  MISCELLANEOUS  DEVICES 

Of  the  many  electrical  devices  used  in  medicine  that  are  not  mentioned  in  the  sections 
above,  three  pieces  of  apparatus  are  briefly  described  in  this  section. 

The  electron  microscope  is  a  device  in  which  a  beam  of  electrons  traverses  a  thin  sample 
of  tissue  or  other  material  and  impinges  upon  a  fluorescent  screen  or  film  to  produce  an 
enlarged  image  representing  a  pattern  of  the  specimen.  The  electron  beam  is  controlled 
by  means  of  electrostatic  or  electromagnetic  lenses.  The  RCA  type  EMU  is  a  recently  de- 
veloped commercially  available  electron  microscope.  The  magnification  can  be  varied 
from  100  to  20,000;  the  resolving  power  is  somewhat  less  than  100  A.  The  device  operates 
with  a  maximum  difference  of  potential  of  50  kvp.  Films  made  on  the  electron  microscope 
can  be  photographically  enlarged  so  that  the  overall  magnification  may  exceed  100,000. 
Among  the  limitations  of  the  device  are:  (1)  it  is  extremely  difficult  properly  to  prepare 
specimens;  and  (2)  the  specimen  may  be  destroyed  or  seriously  modified  in  structure  by 
the  electron  beam. 

The  betatron  is  a  device  for  accelerating  electrons  to  velocities  closely  approaching  the 
speed  of  light.  It  consists  of  a  laminated  iron  core  with  an  air  gap.  Between  the  poles  of 
the  gap  there  is  mounted  a  doughnut-shaped  vacuum  tube  into  which  pulses  of  electrons 
with  a  duration  of  about  2  AISCC  and  velocities  corresponding  to  approximately  80  kvp  can 
be  injected  into  the  evacuated  space.  A  winding  on  the  core  is  fed  from  a  source  of  ISO- 
cycle  alternating  current  in  order  to  produce  across  the  air  gap  a  magnetic  field  in  which 
the  flux  density  varies  sinusoidally  with  time;  the  first  rising  one-quarter  cycle  of  the  flux 
(1/720  sec)  actuates  the  device.  The  pulse  of  electrons  is  injected  into  the  doughnut  a  few 
microseconds  after  the  magnetic  flux  has  passed  through  zero  and  is  increasing.  Sub- 
sequently, the  increasing  magnetic  field  causes  the  electrons  to  be  accelerated;  they  trav- 
erse the  circular  path  within  the  tube  several  hundred  thousand  times  in  1/720  sec.  It  is 
possible  so  to  shape  the  pole  pieces  that  these  electrons  will  remain  in  a  stable  circular 
ofbit  during  their  acceleration.  At  the  end  of  the  period  (1/720  sec)  they  are  caused  to 
spiral  outward  from  the  stable  orbit  and  to  impinge  upon  a  target,  causing  the  production 
of  x-rays.  Betatrons  for  producing  electron  velocities  corresponding  to  20  Mev  have  been 
in  use  for  several  years,  and  an  experimental  model  of  a  100  Mev  betatron  has  been  an- 


HIGH-VOLTAGE   SHOCK  23-17 

nounced  by  the  General  Electric  Company.  At  high  equivalent  voltages,  some  of  the 
phenomena  observed  in  cosmic-ray  studies  have  been  produced  in  the  laboratory  for  the 
first  time. 

A  cyclotron  is  a  device  for  accelerating  protons  and  heavier  positive  ions.  Its  operation 
depends  upon  two  facts:  (1)  charged  particles  traveling  at  right  angles  to  a  constant  mag- 
netic field  traverse  a  circular  path  with  constant  linear  velocity  in  the  absence  of  an  electric 
field ;  (2)  charged  particles  are  accelerated  in  the  presence  of  an  electric  field  in  the  direction 
of  their  motion.  Near  the  center  of  a  shallow,  cylindrical  evacuated  cavity,  positive  ions 
are  emitted  from  a  suitable  source  at  very  low  velocities.  The  cylindrical  space  is  sur- 
rounded by  two  semicircular  dees,  and  the  source  of  ions  is  located  in  the  gap  between  the 
dees.  A  source  of  high-frequency  alternating  voltage  is  applied  to  the  dees  so  that  those 
positive  ions  that  are  emitted  during  the  peak  of  a  particular  half  cycle  are  accelerated 
toward  the  dee  which,  at  that  instant,  is  negatively  charged.  Upon  entering  the  dee  the 
effect  of  the  electrostatic  field  becomes  negligibly  small,  and  the  electrons  would  continue 
in  a  straight  line  except  that  the  dees  are  mounted  in  the  gap  between  the  poles  of  an 
electromagnet  excited  by  direct  current.  Thus  the  ions  that  have  been  initially  ac- 
celerated by  their  first  traverse  of  the  gap  between  the  dees  travel  in  a  semicircle  inside 
the  dees  until  they  again  reach  the  gap.  The  high-frequency  source  of  potential  between 
the  dees  is  adjusted  so  that  the  ion  will  be  again  accelerated  as  it  crosses  the  gap  the  second 
time. 

This  process  is  repeated,  and  the  radius  of  the  semicircle  increases  each  time  the  ions  are 
accelerated  by  the  electric  field.  Thus  the  ions  spiral  outward  and  are  subjected  to  two 
increases  in  acceleration  during  each  revolution.  After  several  hundred  revolutions  the 
ions  spiral  outward  to  the  maximum  diameter  of  the  dees  and  are  then  emitted  tangentially 
through  a  window.  By  placing  various  materials  in  the  path  of  the  beam  outside  the 
cyclotron,  important  nuclear  experiments  can  be  carried  out.  For  example,  when  protons 
are  accelerated  in  the  cyclotron  and  permitted  to  impinge  on  beryllium,  neutrons  are  pro- 
duced in  relatively  large  quantities.  The  largest  of  these  devices,  184  in.  in  diameter,  was 
constructed  at  the  University  of  California  under  the  direction  of  Dr.  E.  O.  Lawrence, 
the  inventor  of  the  apparatus.  The  cyclotron  has  been  used  for  many  important  experi- 
ments in  the  phenomenal  developments  of  nuclear  physics.  Note  that  the  operation  of  the 
cyclotron  in  its  simplest  form  depends  upon  the  following  fact:  The  time  required  for  ions 
to  move  through  one  semicircle  remains  constant  regardless  of  the  linear  velocity  as  long 
as  the  velocity  is  not  comparable  with  the  velocity  of  light.  Ions  in  cyclotrons  with  equiv- 
alent voltages  of  20  Mev  or  less  satisfy  the  condition. 

Several  suggestions  have  been  made  of  methods  for  accelerating  both  electrons  and 
positive  ions  to  velocities  corresponding  to  voltages  of  10s  or  109  volts,  and  development 
of  these  methods  is  now  proceeding. 

BIBLIOGRAPHY 

Electrical  Engineering  Staff  MIT,  Applied  Electronics.     Technology  Press  MIT  and  Wiley  (1943). 

Kerst,  D.  W.,  A  20-million  Electron-volt  Betatron  or  Induction  Accelerator,  Rev.  Scientific  Instrument s, 
Vol.  13,  387-394  (September  1942). 

Lawrence.  E.  0.,  and  M.  S.  Livingston,  Production  of  High-speed  Light  Ions  without  the  Use  of  High 
Voltages,  Phys.  Rev.  (Series  2),  Vol.  40,  19-35  (April  1,  1932). 

Schiff,  L.  L,  Production  of  Particles  beyond  200  Mev,  Rev.  Scientific  Instruments,  Vol.  17,  6-14  (Jan- 
uary 1946). 

HIGH-VOLTAGE  SHOCK  AND  X-RAY  BURN 

19.  HIGH-VOLTAGE  SHOCK 

The  physiological  effects  of  a  high-voltage  electric  shock  may  be  classified  in  two  groups: 
the  major  effects,  such  as  cessation  of  respiration  or  heart  action;  and  the  less  serious  effects, 
such  as  fractures  and  internal  injuries  due  to  falls,  and  also  burns. 

The  first  necessary  action  is  to  remove  the  victim,  from  the  circuit  without  touching  him 
with  bare  hands.  If  it  is  not  possible  to  open  the  circuit  by  means  of  a  switch  near  at  hand, 
then  it  is  usually  effective  to  move  the  conductor,  or  the  victim,  with  a  non-conductor. 
A  physician  should  be  summoned  immediately,  and  it  is  important  to  administer  first  aid 
or  artificial  respiration  immediately  while  awaiting  his  arrival.  If  the  victim  is  breath- 
ing, heart  stimulants  may  be  administered  hypodermically.  The  body  should  be  rubbed  to 
produce  external  warmth,  and  the  clothing  should  be  loosened  in  order  to  excite  con- 
sciousness. 


23-18          MEDICAL  APPLICATIONS  OF  ELECTRICITY 

If  the  victim  does  not  breathe,  artificial  respiration  should  be  applied  as  follows:  Lay 
the  subject  face  down  with  arms  and  legs  extended,  and  turn  the  face  one  side  so  that  the 
mouth  and  nose  are  free  for  breathing.  Remove  foreign  bodies  such  as  tobacco,  gum, 
and  false  teeth  from  the  mouth,  and  have  an  assistant  draw  the  subject's  tongue  forward. 
Kneel,  straddling  the  subject's  thighs,  facing  his  head;  rest  the  palms  of  the  hands  on  the 
muscles  of  the  small  of  the  back  with  fingers  spread  over  the  lowest  ribs.  With  arms  held 
straight,  swing  forward  slowly  so  that  weight  is  gradually  brought  to  bear  upon  the  sub- 
ject. This  operation  should  take  2  or  3  sec.  Immediately  swing  backward,  removing 
the  pressure.  Repeat  this  procedure  12  or  15  times  a  minute,  a  complete  respiration  in 
4  or  5  sec.  Continue  artificial  respiration  at  least  an  hour  without  interruption  or  until 
the  physician  arrives.  Do  not  give  any  liquid  by  mouth  until  the  subject  is  fully  con- 
scious.  After  the  victim  breathes  again,  shock  treatment  may  be  administered  as  out- 
lined above.  If  any  bones  have  been  fractured  or  if  the  victim  appears  to  have  received 
internal  injury  do  not  move  him  any  more  than  is  necessary  and  prepare  for  removal  to 
the  nearest  hospital. 

If  the  victim  has  received  burns  the  surface  of  the  skin  should  be  protected  from  the  air. 
Cut  around  any  clothing  that  sticks  to  the  burns  and  saturate  adhering  cloth  or  cotton 
dressing  with  */2  per  cent  solution  of  picric  acid  or  a  solution  of  baking  soda,  about  1 
teaspoonful  to  a  pint  of  water. 

BIBLIOGRAPHY 

Dalziel,  C,  F.,  and  J,  B.  Lagen,  Effect  of  Electric  Currents  on  Man,  Sleet.  Eng.,  Vol.  60,  63-66  (Feb- 
ruary 1941). 

Ferris,  L,  P.,  B.  G.  King,  P.  W.  Spence,  and  H.  B.  Williams,  Effects  of  Electric  Shock  on  the  Heart, 
£Zec£.#ng.,  Vol.  55,498(1936). 

20.  X-RAY  BURN 

As  noted  under  the  section  on  roentgen  therapy  the  effect  of  x-rays  on  living  tissue  is 
always  destructive.  This  effect  is  cumulative;  that  is,  the  successive  application  of  small 
doses  may  cause  destruction  of  living  tissues.  Workers  in  x-ray  laboratories  should  be 
properly  protected  from  exposure  by  the  installation  of  lead  or  lead  glass  protective  shields 
about  the  x-ray  tube.  If  the  worker  must  be  in  the  field  of  the  x-rays,  he  should  wear  a 
protective  lead  rubber  apron,  hood,  and  gloves.  X-rays  produced  by  low  x-ray-tube 
voltages  are  absorbed  almost  completely  by  the  skin.  To  eliminate  these  x-rays,  which 
have  normally  no  useful  effect  in  treatment  or  roentgenography,  a  filter  of  !/2  mm  or  1 
mm  of  aluminum  is  placed  between  the  x-ray  tube  and  the  patient. 

The  first  effects  of  x-ray  burn  are  reddening  and  itching  of  the  skin  and  falling  hair. 
Later,  open  sores  may  develop,  which  may  subsequently  cause  the  destruction  of  large 
areas. 

Standards  of  x-ray  protection  have  been  set  up  and  internationally  accepted.  These 
standards  should  always  be  observed  rigorously.  It  is  necessary  to  make  periodic  tests  by 
having  workers  carry  small  pieces  of  light-protected  film  with  a  narrow  lead  strip  covering 
part  of  the  outer  casing.  These  films  are  carried  over  a  period  of  several  working  days  and 
are  then  developed  to  discover  whether  any  fogging  has  occurred  on  those  portions  of  the 
film  not  protected  by  the  lead  strip.  Such  fogging  indicates  the  need  of  more  effective 
protective  measures.  lonization  chambers,  similar  in  shape  and  size  to  fountain  pens, 
may  be  charged  and  worn  by  a  worker  for  several  hours,  and  the  exposure  in  fractions  of  a 
roentgen  may  be  measured  by  means  of  a  calibrated  electrometer.  Although  there  is  some 
controversy  concerning  the  magnitude  of  the  safe  daily  tolerance  dose,  a  commonly  ac- 
cepted value  is  0.05  roentgen  per  day. 

BIBLIOGRAPHY 

X-ray  Protection.    Bureau  of  Standards,  Handbook  20. 

Protection  against  X-rays  and  Gamma  Rays,  Radiology,  Vol.  46,  57-76  (January  1946). 
American  War  Standard  Safety  Code  for  the  Industrial  Use  of  X-rays,  Z54. 1-1946,  Am.  Standards 
Assoc.,  New  York. 


INDEX 

NOTE.     The  double  numbers  refer  to  both  section  and  page  numbers.    For  example,  1-38  indicates 
page  38  of  Section  1. 


A  board,  17-08 

A.V.C.,  7-125 

A.W.G.,  1-65,  1-68,  2-10 

Ab,  1-43 

Abamp,  1-46 

Abamperes,  conversion  factors,  1-59 

Abampere-turas,  conversion  factors,  1-64 

Abbreviations  for  engineering  terms,  1-71 

Abcoulombs,  1-43,  1-44,  1-46 

conversion  factors,  1-58,  1-59 
Aberrations,  14-11 

eye,  14-29 
Abfarads,  1-43,  1-46 

conversion  factors,  1-62 
Abhenrys,  1-46 

conversion  factors,  1-63 
Abmhos,  1-46 

conversion  factors,  1-62 
Abohm-centimeters,  conversion  factors,  1-61 
Abohms,  1-46 

conversion  factors,  1-61 
Absolute,  abbreviation,  1-71 

altimeters,  22-32 

coulomb,  1-43  fn 

electrical  system,  1-44 

gain  of  directivity  in  antennas,  6-71 

joule,  1-57 

units  of  electrical  measure,  1-44 
Absorption  coefficients,  acoustical  material,  12-48 
measurement,  12-48 

dielectric,  in  liquids,  2-51 
in  solids,  2-23 

sound,  see  Sound  absorption 
Absorptivity,  12-40 
Abvolts,  1-46 

conversion  factors,  1-60 
A-c  amplifiers,  21-16 

equivalent  circuit  of  vacuum  tubes,  4-07 

-operated  receivers,  filament  power  for,  7-106 

power  used  for  filament  and  plate  supplies  for 

transmitters,  7-108 
Acceleration,  cgs  unit,  1-46 

conversion  tables,  1-53 

linear,  conversion  table,  1-53 

mks  unit,  1-46 

of  gravity,  1-53,  1-79 

symbol  for,  1-46 
Acetone,  2-49 
Acme,  properties,  2-04 
Acoustic  design,  auditoriums,  12-41,  12-69,  12-70, 

12-71,  16-11 

broadcasting  studios,  12-41 
court  rooms,  12-41 
motion-picture  studios,  12-41 
music  rooms,  12-41,  12-74 
theaters,  12-41 ! 

impedance  of  ear,  12-04 

medium,  effects,  13-02 
physical  properties,  13-02 
reaction  of,  on  a  diaphragm,  13-02 

radiators,  13-08 
mechanical  impedance  to  motion,  13-03 


Acoustics,  12-02,  see  also  Sounds 

auditorium,  12-41,  12-69,  12-70,  12-71,  16-11 
decay  curves,  12-49,  12-50 
domed  ceilings,  12-70 
ergodic  state,  of  rooms,  12-41 
geometric,  12-41 

measure  of  reverberation,  12-48 
good,    practical   procedure   for   obtaining,    in 

buildings,  12-76 
procedure  for  obtaining,  12-76 
requirements,  12-40 

polycylindrical  sound  diffusers,  12-70,  12-71 
properties  of  rooms,  12-39 
echo,  12-40 

effective  sound  pressure,  12-39 
intensity  level,  12-40 
mean  free  path,  12-40 
multiple  echo,  12-40 
noise,  12-40 
pressure  level,  12-39 

rate  of  decay,  see  Acoustics,  rate  of  decay 
requirements  for  good,  12-40 
reverberation,  see  Reverberation 
sabin,  12-40 

sound  energy  density,  12-40 
sound  intensity,  12-40 
transmittivity,  12-40 
velocity  level,  12-40 
rate  of  decay,  12-40,  12-48 

curves  obtained  with  high-speed  level  re- 
corder, 12-49 

high-speed  level  recorder,  12-49 
non-linear,  12-49 
ray,  of  rooms,  12-41 
warble  tone  for  reverberation  measurements, 

12-49 

wave,  12-41 

wave  form  produced  at  glottis  in  speech,  12-19, 
12-20 
Acre,  abbreviation  for,  1-71 

-feet,  conversion  factors,  1-49 
Acres,  conversion  table,  1-48 
Acrylates,  2-34 
Acrylic  resins,  2-34 
Active  power,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46,  1-73 
Adaptation,  light,  14-33 
Adcock  antennas,  6-87,  6-88,  22-06 
Address  systems,  16-14 
Adequate  coupling,  6-09 
Adjacent-channel  interference,  in  f-m  systems, 

8-30 

Adjustable  capacitors,  3-55 
inductors,  3-52 
resistors,  3-17 
standards,  11-22,  11-23 
Admittance,  symbol,  1-72 
Admittances,  driving  point,  5-07 
mutual,  5-06 
transfer,  5-07 
Advance,  properties,  2-04 


INDEX 


Aerotronics,  22-30 

Age,  effect  of,  on  hearing,  12-07 

Aging,  3-04 

Air,  2-54 

acoustic  properties,  13-02 

condenser,  3-53,  3-59 

dielectric  constant,  2-54 

dielectric  properties,  2-49 

gap,  2-59,  3-45 

minimum  sparking  potentials,  2-54 

navigation,  radio  aids,  22-04 

facilities  in  new  federal  airways  system, 

22-13 

in  federal  airways  system  today,  22-06 
landing  systems,  proposed,  22-26 
long-range  navigation  systems,  proposed, 

22-31 

miscellaneous,  22-31 
short-range  navigation  systems,  proposed, 

22-28 

terminology,  22-05 
Air-core  inductor,  3-31 

transformers  in  circuits,  6-10 
Airways,  federal  and  civil,  22-02 
Akbar,  properties,  2-04 
Alcomax,  2-66 
Alexanderson  alternators,  7-94 

multiple-tuned  antennas,  6-80 
Alfer,  2-62,  2-70 
Alferon,  properties,  2-04 
Alford  loop  antenna,  6-84 
Algebraic  formulas,  1-02 
Alkyd  resins,  2-34 

Allocation,  television  frequency,  20-20 
Alloy  wires,  of  high  tensile  strength,  2-21 
Alloys,  aluminum-iron,  2-62 

beryllium-copper,  approximate  values  for  the 
physical  properties,  2-11 

cobalt-iron,  2-62 

cobalt-platinum,  2-68 
„  dispersion-hardening,  2-66 

ductile,  2-68 

for  electrical  resistance,  2-03 

Heusler,  2-57 

high-permeability,  2-61 

iron-cobalt,  2-62 

iron-nickel,  2-62 

iron-silicon,  2-61 

properties,  2-04,  2-09 
All-pass  network,  5-21 
All-wave  receivers,  7-124 
Allyl  resins,  2-34 

properties,  2-26 
Allymer,  2-34 
Alnicos,  2-66,  2-67,  2-68 
Alphabet,  Greek,  use  for  symbols,  1-79 
Alsimag,  2-34 

properties,  2-26 

Alternating-current,  abbreviation,  1-71 
Alternators,  7-94 

Alexanderson,  7-94 

Goldschmidt,  7-94 
Altimeters,  absolute,  22-32 
Altunel,  properties,  2-04 
Aluminum  bronze,  properties,  2-04 

-iron  alloys,  2-62 

plates,  1-68 

properties,  2-04,  2-10 

sheets,  1-68 

wire  (tables),  2-17,  2-18 
Alundum,  properties,  2-10 
Alvar,  2-34,  2-47 
A-m  receivers,  f-m  receivers  and,  8-16 

measurement,  11-43 


A-m  receivers,  measurement,  using  standard  test 

loop  antenna,  11-47 
miscellaneous  measurements,  11-50 
Amber,  2-34 

properties,  2-26 

American  Steel  and  Wire  Co.'s  gage,  1-69 
Wire  Gage,  1-66,  1-68,  1-69,  1-70 
Zinc  Gage,  1-66 

Ammeter,  graphical  symbol,  1-76 
Ammonium  dihydrogen  phosphate  (ADP),  13-56 
properties,  13-68 

use  in  piezoelectric  crystals,  13-56 
useful  cuts,  13-69 
Ampere-hour,  abbreviation,  1-71 
-hours,  conversion  factors,  1-58 
-turns,  conversion  factors,  1-64 
Amperes,  1-43,  1-44,  1-45 
abbreviation,  1-71 
conversion  factors,  1-59 
Amphenol  912,  2-34 

Amplification,  see  article  on  particular  device 
Amplification  factor,  4-02,  4-13 
defined,  4-06 
variation,  5-45 

Amplifiers,  7-02,  see  also  Vacuum  tubes 
a-c,  21-16 
audio,  16-06 

of  radio  transmitters,  7-134 
battery  tubes  used  in  combination  for,  7-21 
bidirectional,  7-13 
booster,  16-09 
bridging,  16-27 
broad  i-f ,  7-60 
cascade,  7-03 

cathode  follower,  7-31,  7-47,  7-48 
class  A,  7-02,  7-03 

characteristics,  7-02 

general  use,  7-03 

maximum  plate  power  efficiency,  7-02 

plate  characteristics,  7-07 

summary,  7-15 

tube  characteristics,  7-07 
class  AB,  7-02 
class  AB2,  7-02 
class  B,  7-02,  7-15 

audio,  dynamic  transfer  curves  for  a  1635 
tube,  7-19 

characteristics,  7-02 

distortion,  7-23 

frequency  of  grid  currents,  7-18 

input  resistance,  7-17 

low-power  audio,  7-15 

output  circuit  requirements,  7-18 

plate  efficiency,  7-16 

plate  loss,  7-16 

power-output  calculation,  7-16 

radio,  7-22 

summary,  7-24 

theoretical  maximum  plate  efficiency,  7-02 

type  46,  plate  characteristics,  7-16 
class  C,  7-02,  7-24 

apparent  plate  resistance,  7-26 

characteristics,  7-02 

circuit  calculations,  7-24 

circuit  for  increased  efficiency,  7-26 

efficiency  of  the  plate  circuit,  7-25 

grid-leak  method  of  supplying  bias,  7-25 

plate  efficiency,  7-02 

plate  modulation  and,  7-85 

power  calculations,  7-25 

series  tuned  circuit,  7-88 

summary,  7-27 

typical  circuit,  7-25 
classes,  7-02 


INDEX 


Amplifiers,  combined  shunt  and  series  peaked, 
circuit,  7-37 

plate  load  in  terms  of  frequency  and  total 
capacitance,  7-38 

series  inductance  in  terms  of  frequency  and 
plate  load  resistance,  7-39 

shunt  inductance  in  terms  of  frequency  and 

plate  load  resistance,  7-38 
conditions  for  regeneration,  7-28 
d-c,  21-16 
defined,  7-02 

direct  resistance  coupled,  7-04 
Doherty,  7-131,  7-132 
double-tuned  circuits,  7-64 
effects  of  regeneration,  7-28 
electroacoustic  equipment,  16-14 
grounded-grid,  7-31,  7-49 
harmonics  of,  7-08 
high-power,  7-136 
high-power  audio,  7-22 
i-f,  7-56 

as  source  of  gain  and  selectivity  in  radio  re- 
ceiver, 7-58 

broad,  7-60 

coefficients  of  coupling,  7-59 

critically  coupled  circuits,  7-59 

double-tuned  stage,  7-63 

flat-topped  selectivity  curve,  7-59 

for  a-m  broadcast  receivers,  7-58 

for  f-m  receivers,  7-61 

medium  bandwidth,  7-58 

narrow  bandwidth,  7-58 

of  superheterodyne  receiver,  7-121 

opposing  couplings,  gain  and  bandwidth  of 
i-f  stage,  7-60 

opposing  inductive  and  capacitive  coupling 
and,  7-60 

pulse  technique,  illustrated,  9-14 

tuning  stability,  7-60 

variable  selectivity  of,  7-59 

wide-band,  7-58,  7-63 
in-phase,  7-31,  7-50 

inverse  feedback,  stagger-tuned,  vs.,  7-70 
inverse  feedback  i-f  circuits,  7-68 
klystron,  4-51 

light-weight  class  B  audio,  7-18 
linear,  7-22 

low-plate-resistance  tube,  7-11 
low-power  audio,  7-15 
low-power  intermediate-r-f,  7-136 
modulated,  7-75 
motorboating  of,  7-04 
negative  feedback,  7-31,  7-51 

effect  on  distortion,  7-52 
neutralization  of  grid-plate  capacitance,  7-29 
one-shot,  7-31,  7-53 
oscillator  buffer,  4-52 
parasitic  oscillation,  7-29 
parasitic  self-oscillation,  7-28 
pentode  audio  power,  distortion  in,  7-13 
pentode  power,  7-13 
pentode  voltage,  7-11 
power,  7-131 

and  efficiency  of  grid-bias-modulated,  7-73 

negative  feedback  applied  to,  7-133 

plate-circuit  modulation  used  in,  7-132 

r-f  harmonic  radiation,  7-132 

shunt  neutralization  employed  in,  7-132 
power  amplification,  7-03 
preliminary,  16-06 
prevention  of  oscillation,  7-29 
program,  16-09 
pulse,  see  Pulse  amplifiers 
pushpull,  7-10 


Amplifiers,  radio,  intermediate-r-f,  7-129 
r-c  coupled,  7-04,  7-90 
regeneration  in,  prevention  of,  7-28 
repeaters,  7-15 
response  curve,  7-32 
r-f,  7-22,  7-131 
series  peaked,  7-35 

circuit  for,  7-36 

plate  load  in  terms  of  frequency,  7-36 

series  inductance  in  terms  of  frequency  and 

plate  load,  7-37 
shunt  peaked,  7-33 

circuit  for,  7-33 

plate  load  in  terms  of  frequency,  7-33 

shunt  inductance  in  terms  of  frequency  and 

plate  load,  7-34 
1635  class  B  audio,  driver  and  output  circuit 

for,  7-20 

special-purpose,  7-31 
stagger-tuned,  inverse  feedback  vs.,  7-70 

single-tuned  circuits,  7-64 
suppressor  input,  screen  output,  7-51 
synchronous  single-tuned  circuits,  7-64 
synchronously  tuned,  overall  band  width  of, 

7-64 
transformer-coupled,  7-05 

output  calculations,  7-07 
transformer-coupled  audio,  input  circuit  calcu- 
lations, 7-07 

transformer-coupled  class  A,  7-07 
triode  power,  7-10 
triode  voltage,  7-10 

tube,  performance  calculations  from  tube  con- 
stants, 7-09 

plate  efficiency,  7-09 

power  output,  7-09 
tuned,  7-06 
tuned  coupling,  7-06 
tuned-r-f  receivers,  7-56 
tuned-transformer-coupled,  7-06 
uncompensated,   plate  load  in  terms   of  fre- 
quency, 7-33 

uncompensated  amplifier  stage,  7-32 
unmodulated  intermediate  stages,  7-130 
vacuum  tube,  effect  of  cathode  resistor  and  by- 
pass, 7-44 

variable-gain  pentode  voltage,  7-12 
variation  of  gain  with  frequency  of  resonance, 

6-11 

voltage  amplification,  7-03 
wide-band,  7-31 

alternative  designs,  7-64 

circuit  for,   with  constant-JC  configuration 
low-pass  filter-coupling  network,  7-40 

consisting  of  three  staggered  triples,  7-67 

double-tuned  circuits  used  in,  7-64 

figure  of  merit,  7-64 

formulas,  summary,  7-44 

high-frequency  compensation  methods,  7-43 

high-frequency  response,  7-31 

inverse-feedback  amplifiers  used  in,  7-64 

low-frequency  response,  7-44 

Miller  capacitance  effect  of  tubes,  7-43 

peaking  coil  distributed  capacity,  7-40 

rise  time  of  pulses,  7-64 

stagger-tuned  amplifiers  used  in,  7-64 

synchronous   single-tuned   circuits  used  in, 
7-64 

video  amplifier  response  curves,  7-42 

with    constant-jE-type    coupling    network, 

7-39 
wide-band  i-f,  feedback  pair,  7-68,  7-69 

feedback  triple,  7-68,  7-69 
Amplitude,  distortion,  5-33 


4 


INDEX 


Amplitude,  modulation,  7-71 

distortion  due  to   incomplete  rejection   of, 

8-27 

distortion  from,  8-28 
frequency  modulation  converted  to,  8-17 
incomplete  rejection  of,  f-m  distortion  and, 

8-26 

methods  of  producing,  7-72 
-modulated  radio  transmitters,  7-129 
Amp-turn,  1-46 

Amyl  acetate,  dielectric  properties,  2-49 
Amyl  alcohol,  dielectric  properties,  2-49 
Analysis  of  sound,  11-65 
Analyzers,  commutated  band,  11-66 
'feedback,  11-60 
heterodyne,  11-61 
intermodulation,  11-61 
machine  noise,  11-63 
requirements,  11-56 
resonance,  11-58 
suppression,  11-61 
tuned-reed,  11-65 
Anesthesia,  local,  23-03 
Angle,  cgs  unit,  1-46 
current  flow,  7-131 
hyperbolic,  1-10 
incidence,  5-53 
mks  unit,  1-46 
plane,  conversion  table,  1-51 
reflection,  5-53 
solid,  conversion  table,  1-51 
symbol,  1-46 

trigonometric  functions,  1-07 
Angles  of  rectangular  wave  guides,  10-21,  10-22 
Angstrom  unit,  abbreviation,  1-71 
conversion  factors,  1-47 
international,  1-79 

Angular  acceleration,  conversion  table,  1-53 
frequency,  symbol,  1-73 
localization  of  sounds,  12-18 
velocity,  cgs  unit,  1-46 
conversion  table,  1-53 
mks  unit,  1-46 
symbol,  1-46 
Aniline,  dielectric  properties,  2-49 

formaldehyde,  power  factor  at  high  frequen- 
cies, 2-34 
-formaldehyde  resins,  2-34 

properties,  2-26 
Annealed  copper  standard,  2-20 

defined,  2-02 

Anode  current,  defined,  4-05 
defined,  4-04 
dissipation,  denned,  4-05 
graphical  symbol,  1-77 
voltage,  defined,  4-05 
Anodes,  4-02 
air  cooled,  4-03 
classification,  4-04 
radiation  cooled,  4-03 
water  cooled,  4-03 
Antenna  circuit,  combination  of  capacitative  and 

inductive  coupling,  7-115 
Committee  of  the  Institute  of  Radio  Engineers, 

6-88 

coupling  circuit  of  radio  receiver,  7-115 
coupling  circuits,  7-116 
effect,  defined,  6-87 

sensitivity  of  loop  antenna  to,  6-87 
gain  and  bandwidths,  11-53 
-testing  methods,  study  of,  6-88 
Antennas,  see  also  Radio  antennas 
absolute  gain  of  directivity,  6-71 
absorption,  6-64 


Antennas,  Adcock,  6-87,  6-88,  22-06 
Alexanderson  multiple-tuned,  6-80 
Alford  loop,  6-84 
antenna  images,  6-69 
aperture  of  arrays,  6-73 
area  efficiency  factor  of  apertures,  6-76 
array  formed  by  stacking,  6-73 
arrays,  6-64,  6-73 

for  broadcasting,  6-82 
broadside,  6-72,  6-83 
center  of  radiation,  6-65 
Chireix-Mesney  array,  6-65 
circular,  6-85 
classification,  6-64 
cloverleaf,  6-85 
collimating  devices,  6-78 
conical,  input  impedance,  6-66 
cosecant,  6-87 

of  radar,  6-72 
counterpoise,  6-69,  6-80 
currents  and  voltages  existing  on,  6-66 
curtain  arrays,  6-83 
cylindrical,  input  resistance,  6-67 
cylindrical  optics,  6-77,  6-86 
ctipoles,  electric  and  magnetic,  directional  dia- 
gram of,  6-70 
direction  finding  of,  6-87 
directional,  6-71 

a-m  broadcasting,  16-29 

use  of,  in  medium-frequency  broadcasting, 

6-74 

directional  diagrams  of  isolated  wires,  6-75 
directivity  of,  effect  of  soil  and  terrain,  6-74 
directivity  of  broadcast  transmitting,  6-74 
dummy,  defined,  11-51 
effective  area,  6-72 
end-fire  array,  6-72 

field  intensity  and  loop  in  free  space,  6-70 
field  intensity,  6-70 
fields  associated  with,  6-69 
flat-top,  6-80 
folded  wire  arrays,  6-83 
for  aircraft,  6-88 
for  broadcast  reception,  6-82 
for  high  frequencies,  6-82 
for  low  frequencies,  6-80 
for  medium-frequency  broadcasting,  6-81 
for  very  high  and  ultra  high  frequencies,  6-84 
forms  of,  6-63 
Franklin  arrays,  6-65 
free  space  transmission  law,  6-72 
front  feed  for  reflectors,  6-86 
graphical  symbol,  1-76 
ground  systems  and,  6-69 
grounded-quarter-wave,  6-70 
half-wave,  6-70 

hollow  cylindrical,  input  reactance,  6-67 
horizontal,  directional  diagram  of,  as  influenced 

by  finite  conductivity  of  earth,  6-75 
inverted- V,  6-83 
lenses,  6-78 

line  sources  of  wavelength  power,  6-77 
linear  arrays,  6-64 
linear  conductor,  6-64 

directivity,  6-73 

principles,  6-65 
long-wire,  6-64 
loop,  6-85 

graphical  symbol,  1-76 

sensitivity  to  antenna  effect,  6-87 
magnetic  currents,  6-64 
microwave,  6-86 

primary  feed,  6-77 

radiation,  6-77 


INDEX 


Antennas,  microwave  lobing,  6-87 
Musa  receiving,  6-83 

diversity  reception,  6-83 
non-directional,  6-71 

using  horizontal  polarization,  6-85 
non-dissipative,  aperture,  6-76 

effective  area,  6-76 
omnidirectional,    using   vertical   polarization 

6-84 
optics  of,  cylindrical,  6-77 

spherical,  6-77 
parabolic  reflectors,  6-78 
paraboloid,  beam  width,  6-76 
I      nomogram,  6-76 

•wavelength,  6-76 

point  sources  of  wavelength  power,  6-77 
practical  systems,  6-80 
pylon,  6-86 
quarter-wave,  6-70 
quasi-optical,  6-63 

directivity,  6-76 

lenses,  6-78 

quasi-optical  devices,  6-64 
radiated  power  from  vertical  grounded  wire, 

6-70 

radiation  efficiency,  6-68 
radiation  from,  6-64 
radiation  resistance,  6-68 
RCA  broadside  arrays,  6-65 
rear  feed  for  reflectors,  6-86 
reciprocity,  6-64 
reflectors,  6-78 
resistance,  6-68 
resonant,  6-81 
resonant-V,  6-83 
rhombic,  6-64,  6-83 
rocket,  6-85 

scanning  method,  6-87,  6-88 
shielded  loop,  6-87 
short-wave,  locations,  6-75 
shunt-fed,  6-81 

simple,  components  forming  total  resistance, 
6-68 

equivalent  circuit,  6-67 

vertical  quarter-wave,  6-70 
single  vertical,  directional  diagram,  6-71 
sterba  array,  6-65,  6-82 
tower,  6-81 

transmission  lines  and,  6-88 
transmitting,  a-m  broadcasting,  16-28 

f-m  broadcasting,  16-29 
turnstile,  6-84 
typical  arrays,  6-65 
unidirectional  broadside  arrays,  6-73 
unidirectional  couplet,  6-73 
unzoned  lens  of  reflectors,  6-78 
V-antenna,  6-64 

vertical  grounded  wire,  field  intensity,  6-70 
vertical  half-wave,  directional  diagram  of,  as 
influenced  by  finite  conductivity  of  earth, 
6-75 

voltage  induced  in,  6-69 
wave,  6-64 
wave-guide,  6-64 
zoned  lens  of  reflectors,  6-78 
zoning  of  waves  by  lens,  6-79 
Anticathode,  4-81 
Anti-hyperbolic  sine,  1-10 
Antimony,  properties,  2-04,  2-10 
Anti-sine,  1-08 
Apartments,  noise  levels  acceptable  in,  12-58 

sound  insulation  in,  12-57 
Aperiodic  disturbances,  5-28 
Aperture  distortion,  19-07,  20-30 


Aperture  distortion,  of  arrays,  antennas  and,  6-73 

of  non-dissipative  antenna,  6-76 
Apothecaries'  fluid  measure,  conversion  factors, 
1-50 

weight,  conversion  table,  1-55 
Approximations,  mathematical,  1-16 
Arbitrary  constants,  1-13 
Arc,  conversion  factors,  1-48 

defined,  4-58 

graphical  symbol,  1-76 

resistance,  2-24 
Arc-back,  4-62 

Architect's  measure,  conversion  factors,  1-48   | 
Are,  1-48 
Area,  cgs  unit,  1-46 

conversion  table,  1-48 

efficiency  factor  of  apertures  of  antennas,6-76 

mks  unit,  1-46 

of  segment,  1-18 

symbol,  1-46 

Argentan,  properties,  2-04 
Argon,  dielectric  constant,  2-54 
Arithmetical  progression,  1-02 
Armite,  2-34 
Aroclors,  2-34 

Arrays  of  antennas,  6-64,  6-73 
Arrester,  lightning,  graphical  symbol,  1-76 
Arsenic,  properties,  2-04 
Articulation,  12-27 

clipping,  effects,  12-34 

delay,  effects,  12-36 

extraneous  noise,  effects,  12-33 

frequency  shift,  effects,  12-35 

non-linear  distortion,  effects,  12-34 

percentage,  curves  for  rooms  of  different  sizes 
and  different  times  for  reverberation,  12-74 
for  rooms,  12-69 

phase  distortion,  effects,  12-35 

resonance  type  of  frequency  distortion,  effects, 
12-33,  12-34 

room  reverberation,  effects,  12-38 

tests,  12-31 
Artificial  ear,  12-04 

larynx,  12-21 

respiration,  23-18 

voice,  12-21 
Asbestos,  2-34 

ebony,  2-35 

paper,  2-35 

properties,  2-26 

textiles,  2-35 

wood,  2-35 

Ascoloy,  properties,  2-04 
Askarels,  2-52 
Aspect  ratio,  20-03,  20-20 
Asphalt,  natural,  2-35 

petroleum,  2-35 

properties,  2-26 

relation  between  dielectric  constant  and  resis- 
tivity, 2-51 

sulfurized,  2-35 
Asphaltites,  2-35 
Astigmatism,  14-12 
Atmosphere,  abbreviation,  1-71 

conversion  factors,  1-56 
Atmospheric  noise,  10-42,  10-48 

pressure,  45  deg  cent,  1-79 

normal,  1-79 

Atomic  weight,  abbreviation,  1-71 
Attenuating  band,  of  filters,  6-33 
Attenuation  characteristic  (Tchebycheff  type)  of 
symmetrical  filter,  6-56 

constant,  symbol,  1-72 

equalizer,  16-28 


6 


INDEX 


Attenuation,  measurements,  11-82 

minimum,  5-12,  6-57 

Attenuator  sections  in  passive  circuits,  6-05 
Attenuators,  3-20 

design,  11-99 
Audible  effects  of  phase  distortion,  9-34 

field,  12-05 

frequency  ranges  of  music,  speech,  and  noise, 
12-30 

methods  used  in  frequency  measurement,  11-09 

pressure,  12-05 
Audio  amplifiers,  16-06 

of  radio  transmitters,  7-134 

feedback  factor,  11-48 

-frequency  analysis,  1 1-65 

-frequency  transformers,  see  Transformers,  au- 
dio-frequency 
Audiograms,  23-11 

Auditoriums,  acoustic  design,  12-41,  12-69,  12-70, 
12-71,  16-11 

noise  levels  acceptable  in,  12-58 

sound  level  of  speech  for,  12-73 
Auditory  magnitude  of  sound,  12-11 

nerve  fibers,  12-03 

nerves,  12-02 

excitation  of,  12-03 

conduction  of  neural  pulses  to  brain,  12-04 
pitch  of  low-frequency  tones  and,  12-03 

ossicles,  12-02 

perspective  of  sounds,  12-39 

range,  12-09 

sensation  area,  12-09 
Aural,  radio  range,  22-06 

-visual  range,  22-08 
Austin-Cohen  formula  (sky  wave  propagation), 

10-39 
Automatic  direction  finder,  22-04 

flight,  22-05 

and  landing  equipment,  22-22 

volume  control,  effect  on  the  fidelity  of  receiv- 
ers, 7-125 
A.V.C.,  7-125 
Average,  abbreviation,  1-71 

hearing  loss,  12-08 
Avogadro's  number,  1-79 
Avoirdupois,  abbreviation,  1-71 

weight,  conversion  factors,  1-54,  1-55 
A.W.G.,  1-65,  1-68,  2-10 
Axes,  crystalographic,  13-59 
Ayrton-Perry  coil,  11-19 

B  board,  17-08 

&  S  gage,  1-67,  2-12 

supply  circuit,  7-106 
pl  unit,  1-38 

Babbitt  permeameter,  2-74 
Back  porch,  20-16 

-wall  effect,  15-13 
Bactericidal  radiation,  23-07 
Baffle,  13-11 
Bakelite,  2-35 

Copolymer  Resins,  2-39 

resin,  properties,  2-26 
Balanced  armature  speaker,  13-15 

discriminator,  f-m  distortion  and,  8-28 

power  circuit,  10-77 

response  to  amplitude  modulation,  distortion 

and,  8-28 
Balata,  2-35 
Ballast  lamps,  3-22 

tubes,  4-08 
Balsa  wood,  12-51 
Balsam  wool,  12-53 
Band  frequency  analyzer,  11-60 


Band  speaker,  13-11 
Band-pass  filter,  6-33 

illustrative  design,  6-59 

mechanical,  11-63 
Bandwidth,  5-32 

inadequate,  f-m  distortion  and,  8-26 

television,  20-04,  20-06 
Bank  winding,  3-31 

Banking  rooms,  noise  level  acceptable  in,  12-58 
Bar,  conversion  factors,  1-56 
Barium  titanate,  properties,  2-30 
Barkhausen  effect,  2-69 
Barkhausen-Kurz  Oscillator,  7-91 
Barometer,  abbreviation,  1-71 
Barrel,  abbreviation,  1-71 
Barrier  photocells,  15-13 

frequency  response,  15-15 

illumination  response,  15-13 

sensitivity,  15-14 

structure,  15-13 

wavelength  response,  15-13 
Baryes,  conversion  factors,  1-56 
Base  of  natural  logs,  1-19 
Bass  viol,  power,  12-25 
Battery,  graphical  symbol,  1-76 

tubes,  used  in  combinations  for  amplifiers,  7-21 
Baume,  abbreviation,  1-71 
Beacon,  non-directional,  defined,  22-05 

omnidirectional,  defined,  22-05 
Beads,  20-04 
Beam  coupling  coefficient,  4-53 

transadmittance,  4-53 

transconqluctance,  4-53 

width  of  paraboloid  antenna,  6-76 
Bearing,  defined,  22-05 
Beat  frequency  indicator,  11-05 

frequency  oscillator,  7-89 

methods  used  in  frequency  measurement,  11-06 
Beatnote  interference,  in  f-m  systems,  8-29 
Beeswax,  2-35 

properties,  2-26 
Beetle,  2-35 
Bel,  1-37 

Bellini-Tosi  loop  method  of  direction  finding,  6-88 
Benalite,  2-39 
Bends,  defined,  22-06 

of  rectangular  wave  guides,  10-21,  10-22 
Benzene,  dielectric  properties,  2-49 

relation  between  dielectric  constant  and  resis- 
tivity, 2-51 

Beryllium-copper  alloys,  approximate  values  for 
the  physical  properties  of,  2-1 1 

properties,  2-04 
Bessel  function  in  frequency  modulation,  8-08 

zero,  values  of,  for  first  180  modes  in  circular 

cylinder  resonators,  7-104 
Bessel  functions,  1-37 

chart,  1-37 

in  frequency  modulation,  8-04 

values,  8-05 
BessePs  equation,  1-15 
Betatron,  23-16 
Bias,  4-06 

Bidirectional  amplifier,  7-13 
Binaural  hearing,  16-02 

minimum  audible  sound  field  intensity  levels 
in  hearing,  12-06 

reproduction,  12-39 
Binocular  television,  20-67 

vision,  14-46 
Biotite,  2-39 

"Birdies"  in  intermediate  frequencies,  7-57 
Birmingham  Gage,  1-66 

Wire  Gage,  1-66,  1-69,  1-70 


INDEX 


Bismuth,  properties,  2-04,  2-10 
Black  body,  15-29 

level,  20-13,  20-21 
Blanking,  20-17 
Blocked  impedance,  5-65 
Blocking  oscillators,  9-18 

pulse  duration  and,  9-20 

repetition  rate  of  pulses  determined  by,  9-19 
Blood  pressure,  23-08 
Blurring  interference,  12-70 
Board  measure,  conversion  factors,  1-49 
Boiling  point,  abbreviation,  1-71 
Bolometer,  15-04= 

bridge,  9-11 
circuits,  11-79 

detector,  11-77 

thermistor,  15-04 

Bolometric    methods,    for   low-power   measure- 
ments, 11-77 

Boltzmann  constant,  1-79 
Booster  amplifier,  16-09 
Bootstrap  circuit,  20-10 
Boron,  properties,  2-04,  2-10 
Bougie  decimales,  conversion  factors,  1-65 
Bracket,  square  cross,  5-43 
Brake  horsepower,  abbreviation,  1-71 

-hour,  abbreviation,  1-71 
Brass  plates,  1-68 

properties,  2-04,  2-10 

sheets,  1-68 

Breaking  load,  for  solid  wires,  2-21 
Brick,  12-65 
Bridge  methods,  11-12 

networks,  5-13 
Bridging  amplifier,  16-27 
Briggsian  logarithms,  1-19 
Brightness,  cgs  unit,  1-46 

mks  unit,  1-46 

symbol,  1-46 

television,  20-03,  20-21,  20-66 
Brinell  hardness  number,  abbreviation,  1-71 
British  bushel,  1-50 

gallon,  1-50 

Imperial  pound,  1-42 

Imperial  yard,  1-42 

Standard  Wire  Gage,  1-69,  1-70,  2-14 

thermal  unit,  abbreviation,  1-71 

conversion  factors,  1-57,  1-64,  1-65 
Broadcast  coverage,  range,  10-47 

frequency  allocation,  16-30 

practices,  16-25 

receivers,  11-43 

average  stage  gains  and  second  detector  sen- 
sitivities for  different  types  produced  be- 
tween 1934  and  1946,  7-59 

system,  dynamic  volume  range,  16-34 
layout,  16-26 
tonal  range,  16-33 

Broadcasting,  frequency  modulation,  16-31 
coverage,  16-32 

high-frequency,  16-31 

radio  telephone,   see  Radio  telephone  broad- 
casting 

standard,  16-30 
coverage,  16-32 

station  service,  16-32 
continuity  of,  16-33 
interference,  16-33 

studios,  acoustic  design,  12-41 
sound  insulation,  12-57 

television,  20-21 

transmitter  plant,  16-28 
Broadside  array  antennas,  6-72,  6-83 
Bronze,  properties,  2-04 


Brown  and  Sharpe  wire  gage,  1-66 
Btu,  1-57,  1-65 

Buffer  capacitor,  of  transformers,  6-31 
Buildings,   calculations  of  insulation  in   design, 
12-69 

industrial,  sound  insulation,  12-57 

office,  sound  insulation,  12-57 
Build-up  time,  5-35 
Buna  N,  2-35 

S,  2-35 

Burn,  x-ray,  23-17 
Burrows  permeameter,  2-74 
Bushels,  British  Imperial,  conversion  factors,  1-50 

dry,  conversion  table,  1-49 

heaped,  conversion  factors,  1-50 

struck,  conversion  factors,  1-50 

U.  S.,  conversion  factors,  1-49,  1-50 

II.  S.  Winchester,  1-50 
Busy  test,  17-08 
Butacite,  2-35,  2-47 
Butt  treatment,  10-54 
Butvar,  2-35,  2-47 
Butyl,  2-35 

stearate,  dielectric  properties,  2-49 
Buzz  tone  in  speech,  12-21 
Buzzer  output,  11-65 
BWG,  1-69 

Cable  Morse  Code,  18-04 
Cables,  10-56 

coaxial,  10-57 

conversion  factors,  1-47 

length,  1-47 

loading  coils,  10-58 

sheath  corrosion,  10-63 

principal  causes,  10-63,  10-64 
remedial  measures,  10-65 

sheath  protection,  10-57 

spiral-four  disk-insulated,  10-57 
Cadmium,  properties,  2-04,  2-10 
Calcite,  effective  grating  space,  1-79 
Calcium,  properties,  2-04 

titanate,  properties,  2-30 
Calculus  formulas,  1-12 
Calido,  properties,  2-04,  2-10 
Calorie,  abbreviation,  1-71 
Calorimeter  hi  pulse  measurements,  9-12 
Calorimetric  methods,  for  high-power  measure- 
ments, 11-81 
Calorite,  properties,  2-04 
Camera,  television,  20-23 
Cancer,  23-12 
Candle,  1-43,  1-46 

abbreviation,  1-71 

-hour,  abbreviation,  1-71 

International,  1-43,  1-65 
Candlepower,  abbreviation,  1-71 

mean  horizontal,  abbreviation,  1-72 

spherical,  abbreviation,  1-72 
Candles,  English,  conversion  factors,  1-65 

German,  conversion  factors,  1-65 

International,  conversion  factors,  1-65 
Capacitance,     distributed,     of    audio-frequency 
transformer  winding,  6-24 

electrical,  cgs  unit,  1-46 
conversion  table,  1-62 
mks  unit,  1-46 
symbol,  1-46 

electrode,  denned,  4-06 

electrostatic,  1-72 

formulas,  3-57 

input,  defined,  4-06 

interelectrode,  4-14 
denned,  4-06 


8 


INDEX 


Capacitance,  measurement,  11-24 

audio-frequency  transformers,  6-25 
output,  defined,  4-06 
standards,  11-20 
symbol,  1-72 
variation,  5-41 

Capacitivity,  relative,  symbol,  1-72 
space,  cgs  unit,  1-46 
mks  unit,  l-46f 
symbol,  1-46 
symbol,  1-72 
Capacitors,  3-53 

adjustable,  3-53,  3-55 
air,  plate  spacing,  3-59 
capacitance  formula,  3-57 
ceramic  dielectric,  3-67 
characteristics,  3-54 
classification,  3-53 
effect  of  stray  capacitance,  3-56 
electrolytic,  3-54,  3-68 
fixed,  3-53 

classified  according  to  dielectric  medium,  3-53 
classified  according  to  plate  structure,  3-54 
graphical  symbol,  1-76 
shielded,  graphical  symbol,  1-76 
impregnated-paper,  3-53,  3-60 
metalized  paper,  3-64 
mica,  3-64 
solid-dielectric,  3-53 
variable,  3-53,  3-55 
graphical  symbol,  1-76 
shielded,  symbol,  1-76 
Capacity,  see  Capacitance 
Carat,  conversion  factors,  1-54 
Carbon  dioxide,  dielectric  constant,  2-54 
dielectric  properties,  2-49 
minimum  sparking  potentials,  2-54 
grains,  properties,  2-10 
microphone,  13-26 
monoxide,  dielectric  constant,  2-54 
powder,  properties,  2-10 
properties,  2-04,  2-10 
resistors,  3-11 
steels,  2-66 

tetrachloride,  dielectric  properties,  2-49 
Carbonyl  iron,  2-62 

powder,  2-64 

Carcels,  conversion  factors,  1-65 
Cardiography,  23-08 

Cargo,  measurement  of,  conversion  factors,  1-50 
Carrier,  effect  of  shift,  12-35 
frequencies,  10-04 
telephone  systems,  17-43 
wave  of  modulation,  7-70 
Cascade  amplifiers,  7-03 

cavity,  4-53 
Casein,  2-35 

plastic,  properties,  2-26 
Castor  oil,  2-52 

dielectric  properties,  2-49 

relation  between  dielectric  constant  and  resis- 
tivity, 2-51 
Catalin,  2-35 
Cataphoresis,  23-03 
Catenary,  equation,  1-05 
Cathode  current,  defined,  4-05 

for  long  pulses  in  vacuum  tubes,  9-26 
dark  space,  4-59 
follower  (amplifier),  7-31,  7-47 

line  matching,  7-48 
heating  time,  4-08 
-lead  inductance,  4-16 
•  lenses,  14-58 
-ray  devices,  scanning  circuits,  20-08 


Cathode  -ray  oscilloscopes,  2-76,  11-07 
-ray  tube,  4-02,  15-41,  20-08 
bulbs,  15-43 
defined,  4-04 
displays,  15-49 
for  magnetic   deflection,   graphical  symbol, 

1-77 

oscillograph-type,  15-47 
defined,  4-04 

high-frequency  deflection,  15-47 
scanning  circuits  for,  20-08 
screens,  15-37 

television  picture-reproduction,  15-46 
traces  on,  photography  of,  15-45 
with  electrostatic  deflection,  graphical  sym- 
bol, 1-77 

spot,  defined,  4-75 
Cathodes,  classification,  4-03 
cold,  graphical  symbol,  1-77 
defined,  4-04 
directly  heated,  4-03 

graphical  symbol,  1-77 
filamentary,  4-08 
indirectly  heated,  4-03 

graphical  symbol,  1-77 
ionic-heated,     with     supplementary     heater, 

graphical  symbol,  1-77 
material,  4-03 

photoelectric,  graphical  symbol,  1-77 
pool,  graphical  symbol,  1-78 
thermionic,  4-02 
unipotential,  4-08 

Cathodoluminescence,  defined,  15-30 
Cauterization,  23-03 
Cavity  modulation  in  speech,  12-19 
resonators,  7-95,  11-13 
cavity  couplings,  7-103 
circular  cylinder,  7-97,  7-99,  7-101 
coupling  by  means  of  electron  beam,  7-106 
degeneracy  in  circular  cylinder,  7-101 
design  of  high-Q  cavity  in  TE  Oln  mode, 

7-103 

elliptical  cylinder,  7-101 
formulas,  7-96 

full  coaxial  cylinder,  7-98,  7-101 
mode  chart  for  circular  cylinder,  7-102 
modes,  7-101 
orifice  coupling  of  wave  guide  to  cylindrical, 

7-105 

principle  of  similitude,  7-101 
right  rectangular  cylinder,  7-96,  7-99,  7-101 
Ceiling  isolator,  sound  insulation  and,  12-60 

radio,  10-37 

Ceilings,  domed,  reflection  of  sound  from,  12-70 
Ceilometer,  22-13 
Celeron,  2-36 
Cellophane,  2-35 
Cells,  cesium  oxide,  15-18 

choice  of,  for  various  purposes,  15-15 
photoconductive,  see  Photoconductive  cells 
photoemissive,  see  Photoemissive  cells 
photovoltaic,  15-15 
Celluloid,  2-35,  2-36 

properties,  2-26 
Cellulose  acetate,  2-35 
-butyrate,  2-36 

power  factor  at  high  frequencies,  2-34 
properties,  2-26 
nitrate,  2-36 
properties,  2-26 
propionate,  2-36 

properties,  2-26 

Celotex,  compliance  and  resistance  data  in  insula- 
tion of  vibration,  12-62 


INDEX 


9 


Centiare,  conversion  factors,  1-48 
Centigrade,  conversion  table,  1-64 

units,  1-43 

conversion  table,  1-64 
Centigram,  abbreviation,  1-71 

conversion  factors,  1-54 
Centiliter,  abbreviation,  1-71 

conversion  factors,  1-49 
Centimeter,  1-63 

abbreviation,  1-71 

conversion  table,  1-47 

cubic,  abbreviation,  1-71 
conversion  table,  1-49 

-dynes,  conversion  factors,  1-57 

-gram-second  (system),  abbreviation,  1-71 

-grams,  conversion  factors,  1-57 

square,  abbreviation,  1-72 

conversion  table,  1-48 
Centimetric  waves,  1-SO 
Ceramic  capacitors,  3-67 
Ceramics,  2-36 
Ceresin,  2-36,  2-41 

wax,  properties,  2-26 
Cerex,  2-36 

properties,  2-26 
Cesium  oxide  cells,  15-18 

properties,  2-04 
cgs  electromagnetic  system,  1-43 

electrostatic  system,  1-43 
Chain,  square,  conversion  factors,  1-48 
Chain  (Gunther's),  conversion  factors,  1-47 
Characteristic  (of  logarithm) ,  1-19 

curves,  vacuum  tubes,  4-31 

electrode,  4-06 

impedance,  10-03 

of  speech  sounds,  12-20 
Charge  density,  symbol,  1-72 

electric,  symbol,  1-72 

electronic,  1-79 

line  d.  of,  abbreviation,  1-72 

of  electron,  1-79 

per  unit  area,  conversion  table,  1-59 
Chatterton's  Compound,  2-36 
Chemaco,  2-37 

Chemically  pure,  abbreviation,  1-71 
Chemigum,  2-35 
Chestnut  poles,  10-53 
Cheval-vapeur,  1-58 

China  wood  oil,  relation  between  dielectric  con- 
stant and  resistivity,  2-51 
Chlorinated  diphenyl,  dielectric  properties,  2-49 
Chlorobenzene,  dielectric  properties,  2-49 
Chloroform,  dielectric  properties,  2-49 
Choke  feed,  7-08 

input  filter,  7-108 

swinging,  3-48 
Chromax,  properties,  2-04 
Chrome  steel,  2-63 
Chromel,  properties,  2-04 
Chromium,,  properties,  2-04 
Chronin,  properties,  2-04 
Chronograph,  11-04 
Churches,  acoustic  design,  12-41 

noise  levels  acceptable  in,  12-58 

sound  insulation  in,  12-57 
Cibanite,  2-34 
Cimet,  properties,  2-04 
Circle,  equation,  1-05 

mensuration,  1-18 
Circuit  Q  of  unmodulated  intermediate  amplifier 

stages,  7-130 
Circuits,  see  also  Networks 

air-core  transformers,  6-10 

antenna  coupling,  7-116 


Circuits,  antenna  coupling,  of  radio  receiver,  7-115 
bootstrap,  20-10 
butterfly  tuning,  11-90 
cathode-ray  devices,  20-08 
clamp,  20-16 
clock,  18-40 

coupled,  see  Coupled  circuits 
deflection,  20-08,  20-10 
electric,  5-02 
electronic  control,  21-13 
electronic  power,  21-02 
elements,  6-02 

envelope  delay  distortion  measuring,  11-35 
for  measuring  static  characteristics,  4-08 
impedance  bridge  in,  6-12 
inverse  feedback  i-f  amplifier,  7-68 
linear,  5-37 
measurement,  conductance,  4-12 

electrode  capacitance,  4-14 

Mu  factor,  4-13 

transconductance,  4-12 
neutralizing,  6-12 
non-linear,  5-37 

approximate  series  expansion  for  plate  cur 
rent  of  a  triode,  5-41 

capacitance  variation,  5-41 

characteristics  of  triode  with  load,  5-42 

current-voltage  characteristic,  5-38 

harmonic  analysis  of  current  for  a  sinusoidal 
applied  voltage,  5-46 

inductance  variation,  5-40 

multi-electrode  tubes,  analyses  for,  5-45 

power  series  solution,  5-38 

solution,  5-38 

successive  approximations,  5-45 

trigonometric  series,  5-39 
parallel,  resonance,  6-04 
passive,  attenuators,  6-05 

complex  coupling,  6-07 

elements,  6-02 

pads,  6-05 

phase-measuring,  11-34 
pulse,  see  Pulse  circuits 
pulse  shaping,  see  Pulse  shaping  circuits 
pulse  timing,  see  Pulse  timing  circuits 
receiver,  see  Receiver  circuits 
rectifiers,  7-110 

wave  form  in,  7-107 
regulating,  21-17 
relaxation,  see  Relaxation  circuits 
semi-butterfly,  3-57 
series  resonant,  6-02 

simple  series,  variation  of  current  with  fre- 
quency in,  6-03 

variation  of  voltage  components  with  fre- 
quency in,  6-03 
single-mesh,  5-02,  5-03,  6-02 

circuit  Q,  6-04 

dipoles  in,  6-05 

elements  of  attenuator  sections,  6-05 

matched  impedances  in,  6-06 

non-matched  impedances  in,  6-06 

quadripoles  in,  6-05 

series  resonant,  6-02 

T  section  attenuator  in,  6-06 

T  sections  in,  6-06 

voltage  relations,  6-02 
thyratron  grid-control,  21-19 
timing,  21-14 
transmission,  10-02 

-frequency  characteristic,  6-02 

of  two  single  circuits  in  cascade,  6-10 
transmitter,  for  f-m  transmitters,  8-15 
tuned  amplifier,  6-11 


10 


INDEX 


Circuits,  Wheatstome  bridge,  4-11 

wire  transmission,  see  Wire  transmission  lines 
Circular,  abbreviation,  1-71 

antennas,  6-85 

cylinder  cavity  resonator,  7-97,  7-99,  7-101 
mode  chart,  7-102 

inch,  conversion  factors,  1-48 

mil,  abbreviation,  1-71 
conversion  factors,  1-48 

wave  guides,  10-13 

Clamping  or  d-c  reinsertion  of  pulse  shaping  cir- 
cuits, 9-17,  20-16 
Clapping,  12-30 
Clarinet,  power  of,  12-25 
Class  A  amplifier,  7-02 

B  amplifier,  7-15 

C  amplifier,  7-24 

Classrooms,  noise  levels  acceptable  in,  12-58 
Clearance,  defined,  22-05 
Climatic  loading,  10-51 
Climax,  properties,  2-05 
Clipping,  effect  of,  12-34 
Clock  circuits,  18-40 

synchronons,  11-04 
Close-tolerance  resistors,  3-06 
Cloth,  varnished,  2-47 
Cloverleaf  antennas,  6-85 
Coaxial  cables,  10-57 

lines,  10-05,  10-09  i 

Cobalt-iron  alloys,  2-62 

-platinum  alloys,  2-68 

properties,  2-05 

steels,  2-66 

Co-channel  interference  in  f-m  systems,  8-30 
Cochlea  of  ear,  12-02 

basilar  membrane,  12-03 

canals,  12-03 

scala  media,  12-03 
scala  typani,  12-03 
scala  vestibuli,  12-03 

lamina  spiralis  ossea,  12-03 

nerve,  12-03 

semi-diagrammatic  section,  12-03 
Codes,  color,  3-12 

telegraph,  18-02 
Coefficient,  abbreviation,  1-71 

absorption,  12-48 

decrement,  of  couplings,  6-09 
transmission,  12-63 

coupling,  symbol,  1-72 

resistance-temperature,  symbol,  1-73 
Coercive  force,  defined,  2-59 
Coercivity,  defined,  2-59 
Coil,  3-31 

constant,  10-21 

Cold  cathode,  graphical  symbol,  1-77 
Cold-cathode  tubes,  4-72 

available  types,  4-75 

test,  4-74 

Collimating  devices  of  antennas,  6-78 
C  ©logarithm,  abbreviation,  1-71 
Color  (colors),  14-35 

adaptation,  14-39 

chromaticity  diagram,  14-38 

code,  3-12 

hue  discrimination  threshold,  14-36 

sensation,  14-35 

specification,  14-37 

stimulus,  14-35 

television,. 20-65 

tolerance,  14-39 
Colpitts  oscillator,  7-83 
Coma,  14-12 
Combinations,  1-03 


Comet,  properties,  2-05 
Common  logarithms,  1-19 
abbreviation,  1-71 

year,  conversion  table,  1-51 
Communication  systems,  coordination  of  power 

and,  10-67 
Comol,  2-66,  2-67 
Comparison  measurements,  11-27 
Compatability,  20-66 
Compensated  volume  control,  7-126 
Compensation  theorem,  5-12 
Complementary  impedances,  5-10 
Complex  frequency  plane,  5-04 

notation,  5-03 

quantities,  1-OC 
Compliance,  5-57 

Composition  carbon  resistors,  3-11 
Compound  coupling,  tuned  r-f  tiansformer  em- 
ploying, 6-11 

Compressive  strength,  2-26 
Computers,  flip-flop  circuits,  9-13 

two-digit  decade  counter,  9-09 
Concentrate,  abbreviation,  1-71 
Concentric  tube  line,  10-09 
Concrete,  acoustics  of,  12-66 
Condenser  constant,  11-21 

electrolytic,  3-68 

fixed,  graphical  symbol,  1-76 

input  filter,  7-108 

microphone,  13-24 

speaker,  13-16 

Conductance,  electrical,  cgs  unit,  1-46 
conversion  factors,  1-61 
mks  unit,  1-46 
symbol,  1-46,  1-72 

measurement  circuit,  4-12 

measurement,  11-24 

minimum,  5-08 

of  an  electrode,  4-1 1 
Conducting  materials,  2-02 

definitions,  2-02 

specific,  properties,  2-03 

wire  tables,  2-12 
Conduction,  gaseous,  4-58 

nerve,  see  Nerve  conduction 
Conductivity,  abbreviation,  1-71 

denned,  2-02 

effective,  defined,  2-02 

electrical,  cgs  unit,  1-46 
conversion  table,  1-62 
mks  unit,  1-46 
symbol,  1-46 

equivalent,  symbol,  1-72 

of  gases,  2-54 

of  liquids,  2-50 

symbol,  1-72 

thermal,  conversion  table,  1-65 

units  of,  defined,  2-02 
Conductor,  1-76 
Conduit,  underground,  10-58 
Cone,  right  circular,  mensuration,  1-18 
Conference  rooms,  see  Rooms,  conference 
Confusion,  width  of,  20-04 
Conical  antennas,  input  impedance,  6-66 
Connectors  in  sound  insulation,  12-60,  12-61 
Consonants  of  speech,  12-20 
Constaloy,  properties,  2-05 
Constant  (constants),  abbreviation,  1-71 

arbitrary,  1-13 

attenuation,  symbol,  1-72 

current  modulation,  7-74 

dielectric,  abbreviation,  1-73 

frequency  oscillator,  7-87 

image  transfer,  5-13 


INDEX 


11 


Constant  (constants),  of  integration,  1-13 
physical,  1-79 
time,  symbol,  1-73 
vibration,  symbol,  1-73 
-voltage  generator,  4-07 
wavelength,  symbol,  1-73 
Constantan,  properties,  2-05 
Contact  potential,  4-10 

rectifier,  11-17 
Contactors,  ignitron,  21-11 

thyratron,  21-11 
Continental  Code,  18-03 
Continuity  equation,  5-50 
Continuous  waves,  9-02 
Contours  of  loudness,  12-15 
Contrast,  roentgenographic,  23-14 

television,  20-03 
Control  electrode,  4-04 
grid,  4-04 

-grid  plate  transconductance,  4-06 
noise,  10-82 

Conversation,  power  of,  12-22 
Conversion  factors,  1-42 
gain,  11-53 
tables,  1-47 

angular  acceleration,  1-53 

angular  velocity,  1-53 

area,  1-48 

capacitance,  1-62 

charge  per  unit  area  and  electric  flux  density, 

1-59 

current  density,  1-59 
density  or  mass  per  unit  volume,  1-55 
electric  conductivity,  1-62 
electric  current,  1-59 
electric  field  intensity  and  potential  gradient, 

1-60 
electric  potential  and  electromotive  force, 

1-60 

electric  resistance,  1-61 
electric  resistivity,  1-61 
energy,  work,  and  heat,  1-57 
force,  1-55 
inductance,  1-63 
length,  1-47 
light,  1-65 

linear  accleration,  1-53 
linear  velocity,  1-52 
magnetic  field  intensity,  potential  gradient, 

and  magnetizing  force,  1-64 
magnetic  flux,  1-62 

density,  1-62 
magnetic  potential  and  magnetomotive  force, 

1-64 

mass  and  weight,  1-54 
plane  angle,  1-51 

power  or  rate  of  doing  work,  1-58 
pressure  or  force  per  unit  area,  1-56 
quantity  of  electricity  and  dielectric  flux, 

1-58 

solid  angle,  1-51 
specific  heat,  1-64 

thermal  conductivity  and  resistivity,  1-65 
time,  1-51 

torque  or  movement  of  force,  1-56 
volume,  1-49 
transconductance,  4-22 

defined,  4-06 
Coordination,  induction,  10-73 

structural,  10-68 
Copaline,  2-36 
Copel,  properties,  2-05 
Copper  chloride,  properties,  2-10 
-clad  steel  wire,  2-20 


Copper-iron,  properties,  2-05 
-manganese,  properties,  2-05 
-manganese-iron,  properties,  2-05 
-manganese-nickel,  properties,  2-05 
oxide,  properties,  2-10 
plates,  1-68 
properties,  2-03,  2-10 
sheets,  1-68 
standard,  2-02 

temperature  coefficients,  2-03 
wire  (tables),  2-12,  2-13,  2-14,  2-15,  2-16 
Cord,  abbreviation,  1-71 

of  wood,  conversion  factors,  1-49 
Cordierite,  2-36 

ceramic,  properties,  2-26 
Core  loss,  2-59 

of  power  transformers,  6-28,  6-29 
resistance,  measurements  of,  audio-frequency 

transformer,  6-25 

materials  for  pulse  applications,  9-28 
Corkboard,  compliance  and  resistance  data  in  in- 
sulation of  vibration,  12-62 
Co-ro-lite,  2-36 
Corona,  2-54 
Corprene,  2-36 
Corronil,  properties,  2-05 
Cos,  1-07,  1-21 
Cosecant,  1-07 
abbreviation,  1-71 
antennas,  6-72,  6-87 
hyperbolic,  1-10 
Cosh,  1-10,  1-26 
Cosine,  1-07 

abbreviation,  1-71 
hyperbolic,  1-10 

abbreviation,  1-71 
Cot,  1-07 
Cotangent,  1-07 
abbreviation,  1-71 
hyperbolic,  1-10 
Coth,  1-10,  1-26 

Cottonseed  oil,  relation  between  dielectric  con- 
stant and  resistivity,  2-51 
Coulombs,  1-44,  1-45,  1-46 
absolute,  1-43  fn 
conversion  factors,  1-58,  1-59 
Coulomb's  law,  1-46 

Counter  electromotive  force,  abbreviation,  1-71 
Counterpoise,  6-69,  6-80 
graphical  symbol,  1-76 
Counters,  frequency,  9-13 
Coupled  circuits,  6-02,  6-06 
audio-frequency  transformers,  6-13 
common  impedance  in,  6-06 
complex  coupling,  6-07 
currents,  6-07 
mutual  impedance,  6-06 
overall  selectivity  curve,  6-10 
pure  coupling,  6-07 
resonant,  staggered  tuning,  6-10 
selectivity,  6-10 
self-reactances,  6-07 
self-resistances,  6-07 

transmission-frequency  characteristic,  6-09 
voltages,  6-07 

Couplet,  unidirectional,  6-73 
Coupling  factor,  6-07 

time  constant  of  pulse  amplifiers,  9-14 
units,  flexible,  10-18 
Couplings,  adequate,  6-09 
capacitance,  6-07 
coeflicients,  6-09,  6-10 

symbol,  1-72 
combined  self  and  mutual  inductance,  6-07 


12 


INDEX 


Couplings,  compound,  tuned  r-f  transformer  em- 
ploying, 6-11 
critical,  6-08 
denned,  6-08 
selectivity,  6-10 
decrement  coefficients,  6-09 
deficient,  6-09 
degrees,  6-09 

effect  of  leakage  reactance,  6-11 
inadequate,  6-09 
inductance,  6-07 
loop,  graphical  symbol,  1-77 

output,  4-47 
mutual  inductance,  6-07 
output,  magnetrons,  4-47 
reactance,  6-07,  6-08,  6-09 
resistance,  6-07,  6-08,  6-09 
selectivity  of  i-f  amplifiers  and,  7-59 
sub,  6-09 
sufficient,  6-09 
super,  6-09 
types,  6-07 

wave-guide  output,  4-47,  4-48 
Course  sharpness,  denned,  22-05 
Court  rooms,  acoustic  design,  12-41 
Coverage,  police  radio,  16-37 
Critical  angle,  5-54 
coupling,  6-08 
defined,  6-08 
selectivity,  6-10 
grid  potential,  4-59 
voltage,  4-40,  4-59 
Cronin,  properties,  2-05 
Cronit,  properties,  2-05 
Crookes  tube,  4-58 

Crosby  method  of  frequency  control,  8-13 
Cross  arm,  10-55 
lime,  11-41 
modulation,  4-24 
Crossbar  telephone  system,  17-26 
crossbar  switch,  17-26 
marker,  17-26 
multi  contact  relay,  17-28 
operation,  17-29 
Crosstalk,  far-end,  11-38 
interference,  in  f-m  systems,  8-29 
measurement,  11-37 
near-end,  11-38 
Crystal  detector,  7-76 

graphical  symbol,  1-76 
receivers,  7-117 
oscillators,  7-92 
Crystals,  axes,  13-56 
ferroelectric,  13-57 
microphone,  13-25 

piezoelectric,  see  Piezoelectric  crystals 
Csc,  1-08 
Csch,  1-10,  1-26 
Cubic,  abbreviation,  1-71 
centimeter,  abbreviation,  1-71 

conversion  table,  1-49 
equations,  1-03 

feet,  conversion  table,  1-49,  1-50 
per  minute,  abbreviation,  1-71 
foot,  abbreviation,  1-71 
inch,  abbreviation,  1-71 

conversion  table,  1-49,  1-50 
measure,  1-49 
meter,  abbreviation,  1-71 
conversion  table,  1-49 
yard,  abbreviation,  1-71 
conversion  table,  1-49 
Cumene,  dielectric  properties,  2-49 
Cunico,  2-67,  2-68 


Cunife,  2-67,  2-68 
Cupron,  properties,  2-05 
Cuprous  oxide  cell,  15-13 
Curie  point,  2-70 

Current  (currents),  anode,  defined,  4-05 
cathode,  denned,  4-05 
density,  cgs  unit,  1-46 
conversion  table,  1-59 
mks  unit,  1-46 
symbol,  1-46,  1-72 
electric,  cgs  unit,  1-461 
conversion  table,  1-59 
mks  unit,  1-46 
symbol,  1-46 
filament,  4-08 

defined,  4-05 
gas,  defined,  4-05 
grid,  defined,  4-05 
heater,  4-08 

defined,  4-05 
in  coupled  circuits,  6-07 
ionization,  4-10 
leakage,  4-10 

defined,  4-05 
loops,  5-26 
maximum  secondary,  conditions  for  and  value; 

of,  in  two  mesh  circuits,  6-09 
measurement,  11-16 
nodes,  5-26 
non-sinusoidal,  5-02 
of  antennas,  6-66 
plate,  denned,  4-05 
space-charge  limited,  4-02 
symbol,  1-72 

temperature  limited,  4-02 
transferred  reactance,  6-08 
transferred  resistance,  6-08 
Curtis  winding,  11-19 
Curve  (curves),  common,  equations,  1-05 
demagnetization,  2-59 

of  permanent-magnet  materials,  2-65 
grid  characteristic,  4-10 
magnetization,  2-68 
permeability,  2-57 
universal  resonance,  6-04 
Cutoff  frequencies,  of  filters,  6-33 

of  networks,  6-38 
parabola,  4-40 
voltage,  4-40 

formula,  4-40 

Cycles  per  second,  abbreviation,  1-71 
Cyclohexane,  dielectric  properties,  2-49 
Cycloid,  mensuration,  1-17 
Cyclotron,  23-17 

frequency  magnetrons,  4-41 
Cylinder,  right  circular,  mensuration,  1-18 
Cylindrical  optics  of  antennas,  6-77,  6-86 
Cymbals,  power  of,  12-25 
Cymene,  dielectric  properties,  2-49 

Damped  impedance,  5-65 

wave  transmitter,  7-94 
Damping  constant  or  coefficient,  symbol,  1-72 

constants  of  sounds,  12-20 
Davisson  coordinates,  4-09 
Days,  conversion  table,  1-51 
db,  1-38 
dbm,  1-41 
dby,  1-38 
D-c  amplifiers,  21-16 

and  low-frequency  line  testing,  11-41 

component,  television,  20-03 

control  power  supplies,  stabili/ed,  21-14 

reinsertion,  9-17,  20-16 


INDEX 


13 


D-c  telegraph  system,  18-18 
Dead-end  filter,  5-32 
Deafness,  12-07 

visible  hearing  and,  12-21 

visual  telephony  and,  12-21 
Decade  resistance  boxes,  3-20 

standard,  11-20 

Decahydronaphthalene,  dielectric  properties,  2-49 
Decane,  dielectric  properties,  2-49 
Decay  of  sound,  12-42 

curves,  12-46 
Decibels,  1-37 

abbreviation,  1-71 
Decigrams,  conversion  factors,  1-54 
Deciliter,  conversion  factors,  1-49 
Decimal  Gage,  1-66 

Decimales,  bougie,  conversion  factors,  1-65 
Decimeters,  conversion  factors,  1-47 
Decimetric  waves,  1-80 
Decineper,  1-37 
Decrement,  6-04 

Decylene,  dielectric  properties,  2-49 
De-emphasis  low-pass  filter,  f-m  receivers  and, 

8-31 

Deficient  coupling,  6-09 
Definition,  television,  20-02 
Deflection,  20-08 

electrostatic,  20-08 

output  amplifier,  20-61 

magnetic,  20-10 

output  amplifier,  20-61 

sensitivity,  15-42 
Degree,  abbreviation,  1-71 

Centigrade,  abbreviation,  1-71 

conversion  table,  1-51 

Fahrenheit,  abbreviation,  1-71 

Kelvin,  abbreviation,  1-71 

of  freedom,  5-56 

Reaumur,  abbreviation,  1-71 
Deionization  time  of  gas-filled  oscillators,  7-91 
Dekagram,  conversion  factors,  1-54 
Dekaliter,  conversion  factors,  1-49 
Dekameter,  conversion  factors,  1-47 
Delay,  5-36 

characteristics  in  f-rn  distortion,  8-26 

defined,  5-36 

distortion,  5-16 

envelope,  5-36 

intercept,  5-36 

lines  for  pulse  circuits,  9-28 

phase,  5-36 

sound,  12-70 
Delta  function,  5-27 
Demagnetization  curve,  2-59 

of  permanent-magnet  materials,  2-65 
Demagnetizing  factor,  2-73 
Demodulation,  defined,  7-76 
Density,  cgs  unit,  1-46 

charge,  abbreviation,  1-72 

conductors,  2-04 

conversion  table,  1-55 

current,  symbol,  1-72 

energy,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 

flux,  magnetic,  cgs  unit,  1-46 
conversion  table,  1-63 
mks  unit,  1-46 
symbol,  1-46 

mks  unit,  1-46 

roentgenographic,  23-14 

symbol,  1-46 

Deposited-carbon  resistors,  3-15 
Depth,  localization,  12-18 


Depth,  perception  of,  14-45 

Derivatives,  1-12 

Design,  building,  calculation  of  insulation,  12-69 

transformers,  6-22,  6-23,  6-28,  6-31 
Destruction  of  tissue,  23-12 
Detection,  crystals  and,  7-80 
denned,  7-76 
distortion-free,  7-81 
linear,  7-79 

detection  mutual  conductance,  7-81 
detection  plate  resistance,  7-81 
transrectification  factor,  7-81 
rectification  diagrams  of  detecting  device,  7-79 
square-law,  7-76,  7-78 

detector  circuits,  7-78 
static  characteristic  of  diode,  7-77 
static  characteristic  of  iron  contact  on  ferro- 

silicon,  7-77 

static  characteristic  of  triode,  7-77 
vacuum  tubes  and,  7-80 
Detectors,  7-76,  11-31 

crystal,  graphical  symbol,  1-76 

devices  serving  as,  7-76 

diode,  7-79 

distortion  and,  7-80 

frequency,  see  Frequency  detectors 

grid  current,  7-79 

ideal  characteristic,  7-80 

load  rectification  diagram  for  diode,  7-81 

peak,  7-79 

plate  current,  resistance  coupled  to  succeeding 

amplifier  tube,  7-78 
pulse,  9-24,  9-26 
sensitivity,  denned,  11-50 
square-law,  detection  of  carriers  and,  7-76 

single  sideband  signals  and,  7-76 
tetrode,  7-82 

load  rectification  diagrams,  7-82 
load  rectification  of,  7-82 
Determinants,  1-04 

Deviation  ratio  of  frequency  modulation,  8-03 
Devices,  electro-optical,  15-02 

photoresponsive,  see  Photoresponsive  devices 
thermal,  see  Photoresponsive  devices,  thermal; 

Thermal  devices 
DF  stations,  22-04 
Diamagnetic  materials,  2-57 
Diameter,  abbreviation,  1-71 
Diaphragms,  13-08 
cone,  13-09 
curvilinear,  13-09 
elliptical,  13-09 
Diathermy,  apparatus,  23-04 

technique,  23-06 
Diatonic  scale,  11-09 
Dichlorodifluoromethane,  2-56 
Dielectric  absorption,  in  liquids,  2-51 

in  solids,  2-23 

coefficient  for  free  space,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 
constant,  of  gases,  2-53 
of  insulating  material,  2-21 
of  liquids,  2-48 
of  solids,  2-29 
symbol,  1-73 

flux,  conversion  table,  1-58 
lenses,  use  in  antennas,  6-78 
losses,  4-18 

materials,  solid,  2-25    . 
properties,  of  insulating  materials,  2-21 
strength,  of  gases,  2-54 
of  liquids,  2-51 
of  solids,  2-23 


14 


INDEX 


Dielectrics,  gases  as,  2-53 

liquid,  2-48 
Difference  limen  of  hearing,  12-09 

of  potential,  1-73 
Differential  equations,  1-13] 

operator,  5-29 

sensitivity  of  hearing,  12-09 
Differentiation  of  pulse  in  pulse  shaping  circuits, 

9-15 

Diffraction,  10-32 
Dilectene,  2-34,  2-36 
Dilecto,  2-36 
Dilver,  properties,  2-05] 
Diode,  4-02 

defined,  4-04 

detector,  7-79 

gas-filled,  4-02 

graphical  symbol,  1-77 

load  impedance,  ratio  of  alternating-current  to 
direct-current  in  i-f  amplifiers,  7-59 

pentode  tube  of  superheterodyne  receiver,  7-121 
Diplexing  of  picture  and  sound,  20-67 
Diplopia,  14-28 
Dipole,  6-62 
Direct  capacitance,  11-24 

current,  abbreviation,  1-71 

frequency  modulation,  8-12 
frequency  control  of,  8-13 

radiator  speaker,  13-11 

resistance  coupled  amplifiers,  7-04 
Direction  finders,  airborne,  22-05 
automatic,  22-04 

finding,  22-04 
_     of  antennas,  6-87 

of  rotation  (trigonometric),  1-06 
Directional  baffle,  13-11 

characteristics  of  long  wires  in  antennas,  6-74 

couplers,  11-73 

diagrams  of  horizontal  and  vertical  half-wave 

antennas,  6-75 
Directivity,  of  hearing,  12-07 

of  linear  conductor  antennas,  6-73 

of  quasi-optical  antennas,  6-76 

of  quasi-optical  horns,  6-76 

principle  of,  antennas  and,  6-71 
Discrete  frequency  analysis,  11-59 
Discrimination  ratio,  7-115 
Discriminators  as  frequency  detectors,  8-19 
Dispersion-hardening  alloys,  2-66 
Displacement,  electrical,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46,  1-73 
Display  devices,  20-08 
Dissector,  image,  see  Image  dissector 
Dissipation  factor,  2-22 
Distance  measurement,  9-09 
reflection  of  pulse,  9-10 
return  of  pulse,  9-10 

-measuring  equipment,  22-05,  22-20 

range,  defined,  22-06 

viewing,  20-02 
Distortion,  5-16,  14-12 

amplitude,  5-33 

correctors  in  passive  circuits,  6-05 

delay,  5-16 

effects  on  speech  and  music,  12-29 

-free  detection,  7-81 

frequency,  5-16 

effect  on  sounds,  12-30 
modulation,  see  F-m  distortion 

in  driver  transformer,  6-22 

in  f-m  systems,  8-26 

intermodulation,  measurement,  11-33 

introduced  by  r-f  amplifier  tubes,  4-24 


Distortion,  linear  passive  networks,  5-16 

modulation,  4-24 

non-linear,  5-38 

effects  on  articulation,  12-34 

phase,  5-33,  20-05 
audible  effects,  12-37 

second  harmonic,  4-24 

telegraphy,  18-11 

teletypewriter,  18-15 

third  harmonic,  4-24 
Distortionless  lines,  5-26 
Distributed  capacitance,  3-03,  3-32,  6-14 
Diurnal  characteristic,  10-39,  10-41 
Diverse  waves  in  distance  observation,  9-09 
Diversity  reception  of  Musa  receiving  antenna, 

6-83 

Dividers,  frequency,  9-13 
Dodecane,  dielectric  properties,  2-49 
Doherty  amplifier,  7-131,  7-132 
Domains,  magnetic,  2-68 
Doors,  coefficients  of  sound  transmission,  12-65 

noise-reduction  factor  and,  12-69 
Dosage,  x-ray,  23-13 
Dot  sequential  television,  20-65 
Dozen,  abbreviation,  1-71 
Double-pulse  generator,  9-24 

-tuned  circuits  of  amplifiers,  7-64 

-tuned  transformers,  three  cascade,  f-m  distor- 
tion from,  8-27 
two  cascade,  amplitude  characteristic,  8-28 

vibration,  11-02 
Doublet,  5-52,  6-63 
Downward  amplitude  modulation,  f-m  distortion 

and,  8-29 

Drachm,  conversion  factors,  1-50,  1-54,  1-55 
Dram,  abbreviation,  1-71 
Dri-film,  2-36,  2-45 
Drift  space,  4-51 
Drive-in  theaters,  16-25 
Driver  circuits,  7-22 
Driver  transformers,  6-22 

distortion,  6-22 

frequency  response,  6-22 

leakage  reactance,  6-22 

turns  ratio,  6-22 

winding  arrangement  of  class  B,  6-22 
Driving  point  admittance,  5-07 

impedance,  5-06,  5-10 
Drum,  power  of,  12-25 
Dry  measure,  conversion  factors,  1-50 
Duality,  principle,  5-07 
Ductile  alloys,  2-68 
Dumet,  properties,  2-05 
Dummy  antenna  loads,  3-22 
Durez,  2-36 
Durite,  2-36 

DV  (double  vibration),  11-02 
Dynamic  plate  resistance,  5-45 

speaker,  13-11 

transfer  characteristic,  7-15 
Dynatron  oscillator,  7-89 
Dyne,  1-46 

Dyne-centimeters,  conversion  factors,  1-56 
Dynes,  conversion  factors,  1-55 
Dynode,  graphical  symbol,  1-77 

E  layer,  10-38 

€  (base  of  natural  logs),  1-19 

Ear,  acoustic  impedance,  12-04,  12-05 

as  viewed  through  aperture  of  receiver  cap, 

12-05 

artificial,  12-04 

auditory  nerves,  see  Auditory  nerves 
auditory  ossicles,  12-02 


INDEX 


15 


Ear,  canal,  12-02 
cochlea,  12-02 
description,  12-02 
differential  sensitivity  and,  12-09 
eustachian  tube,  12-02 
inner,  12-03 

helicotrema,  12-03 
semicircular  canals,  12-03 
stapes  attached  to,  12-02 
stirrup,  12-02 

left,  semi-schematic  section,  12-02 
middle,  12-02 
hammer,  12-02 
malleus,  12-02 

natural  frequency  and  damping  constant,  12-04 
outer,  12-02 
oval  window,  12-02 
pinna,  12-02 
round  window,  12-02 
semicircular  canal,  12-02 
sensitivity,  12-05,  12-06 
auditory  range,  12-09 
auditory  sensation  area,  12-09 
classes  of  determinations,  12-05 
difference  between  better  ear  and  average  of 

both  ears,  12-06 
difference  limen  and,  12-09 
frequency  difference  limen  and,  12-10 
intensity  difference  limen  and,  12-10 
minimum  audible  field,  12-05 
minimum  audible  pressure,  12-05 
monaural  minimum  pressure,  12-07 
population  and,  12-06 
possible  lower  limits,  12-06 
pressure  levels  and,  12-05 
sound  intensity  and,  12-05 
stimulation  density  of  auditory  nerve  endings 

and,  12-12 

thermal-acoustic  noise  and,  12-06 
threshold  of  feeling,  12-09 
threshold  of  hearing,  12-05 
zero  hearing  loss,  12-07 
Eardrum,  12-02 

malleus  attached  to,  12-02 
Earphones,  see  Telephone  receivers 
Earth  resistivity,  effect  of,  10-89 
Ebonite,  2-36 
EC,  2-37 

Echelette  grating,  11-13 
Echo,  12-40,  12-70 
flutter,  12-40 
in  auditoriums,  12-70 
multiple,  12-40 
testing  of  lines,  11-38 
Eddy-current  loss,  2-59 
Edison  Wire  Gage,  1-69 

Effective  area  of  non-dissipative  antennas,  6-76 
conductivity,  defined,  2-02 
height,  6-70 
inductance,  3-03 
plate  resistance,  5-45 
resistance,  measurement,  11-27 
spot  width,  20-04 
values,  1-74 
Efficiency,  abbreviation,  1-71 

of  audio-frequency  transformers,  6-16 
of  output  transformers,  6-18 
plate,  7-09 
symbol,  1-73 

Einthoven  galvanometer,  23-09 
Elastance,  symbol,  1-73 
Elastivity,  symbol,  1-73 
Electric,  abbreviation,  1-71 
charge,  symbol,  1-72 


Electric,  circuits,  see  Circuits 
conductance,  1-61 
conductivity,  1-62 
displacement,  symbol,  1-73 
doublet,  5-52 

field  intensity,  cgs  unit,  1-46 
conversion  table,  1-60 
mks  unit,  1-46 
symbol,  1-46 
flux,  cgs  unit,  1-46 

density,  conversion  table,  1-59 
rnks  unit,  1-46 
symbol,  1-46 
induction,  10-73 
intensity,  symbol,  1-73 
moment,  symbol,  1-73 
polarization,  symbol,  1-73 
potential,  conversion  table,  1-60 
resistance,  1-61 
resistivity,  1-61 
units,  1-43 

practical,  1-44 
wave  filters,  see  Wave  filters 
Electrical  conductance,  1-61 
equivalent  of  heat,  1-79 
quantities,  measurement,  11-16 
systems,  5-56 

comparison  with  mechanical,  5-59 
energy,  5-57 
units,  1-43 
Electricity,  medical  applications,  23-02 

quantity  of,  symbol,  1-73 
Electris,  properties,  2-05 
Electroacoustic  devices,  aids  to  the  deaf,  23-11 
equipment,  16-11 
amplifiers,  16-14 
control  equipment,  16-14 
transducer,  13-02 

Electrocardiography,  apparatus,  23-08 
Electrochemical  cauterization,  23-03 
Electrode  admittance,  defined,  4-05 
capacitance,  defined,  4-06 

measurement  circuit,  4-14 
characteristic,  defined,  4-06 
conductance,  defined,  4-05 
deflecting,    reflecting    or    repelling,    grapnical 

symbol,  1-77 
impedance,  defined,  4-05 
resistance,  4-11 
defined,  4-05 
Electrodes,  conductance,  4-11 

control,  defined,  4-04 
Electrodynamic  speaker,  13-11 
Electroencephalography,  apparatus,  23-08 
Electrolysis,  10-63 
Electrolytic  capacitors,  3-68 
Electromagnetic  coupling,  5-64 
horn  (antenna),  6-79 
radiation,  5-49 

fields  due  to  a  current  in  a  wave,  5-52 
Maxwell's  equations,  5-50 
progressive  plane  waves,  5-51 
reflection,  5-52 
refraction,  5-52 

Electromechanical-acoustic  devices,  13-02 
-acoustic  systems,  5-62 
analogues,  5-60,  5-63 
frequency  control,  8-14 
oscillators,  7-91 
systems,  5-56,  5-64 

electrostatically  coupled,  5-62 
interaction  factors,  5-65 
magnetically  coupled,  5-64 
stabilization,  21-32 


16 


INDEX 


Electromotive  force,  abbreviation,  1-71 

cgs  unit,  1-46 

conversion  table,  1-60 

measurement,  3-02 

mks  unit,  1-46 

physiological,  23-08 

symbol,  1-46,  1-73 

unit,  1-44 
Electron  currents,  stray,  4-11 

emission,  4-10 
defined,  4-04 

gun,  15-41 

operation,  15-42 

mass  of,  1-79 

microscope,  23-16 

optical  systems,  general  theorems  on,  14-63 

optics,  14-49 

prisms,  14-62 

transit  angle,  4-53 

tubes,  Section  4;  see  also  Gaseous  conduction 
tubes;    Klystrons;   Magnetrons;  Vacuum 
tubes ;  X-ray  tubes 
Electronic  charge,  1-79 
symbol,  1-73 

control  equipment,  21-02 

controls,  d-c  motor,  21-27 

process  instrumentation,  21-32 
resistance-welder,  21-23 
side-register  positioning,  21-30 

devices,  21-20 

numerical   integrator   and   computer    (eniac), 
9-08 

relays,  contact-operated,  21-20 

tuning,  4-55 

sink  margin,  4-56 
Electrons,  mass,  4-14 

transit  time,  4-15 

velocity,  4-14 

Electro-optical  devices,  15-02 
Electrose,  properties,  2-26 
Electrostatic  coupling,  5-62 

deflection,  output  amplifiers,  20-61 
scanning  circuits,  20-08 

lenses,  14-51 

system  of  units,  1-43 
Electrosurgery,  23-06 
Electrotherapy,  apparatus,  23-02 

electrochemical  cauterization,  23-03 

galvanic  current,  23-03 

iontophoresis,  23-03 

kataphoresis,  23-03 

miscellaneous,  23-03 
Elinvar,  properties,  2-05 
Ellipse,  equation,  1-05 

mensuration,  1-18 
Ellipsoid,  mensuration,  1-18 
Elliptical  cylinder  cavity  resonators,  7-101 
Emergency  transmitters,  frequency  modulation, 
8-15 

phase  modulators  used  for,  8-15 
Emission  characteristic,  4-09 

defined,  4-05 
Emissions,  electron,  4-10 

grid,  4-11 
Empire,  2-36 
Enamel,  2-36 

varnish,  2-36 

vitreous,  2-36 
properties,  2-26 

wire,  2-36 

End-fire  array  antennas,  6-72 
Energy,  cgs  unit,  1-46 

conversion  table,  1-57 

density,  cgs  unit,  1-46 


Energy,  density,  mks  unit,  1-46 
symbol,  1-46 

dissipation,  resistors,  3-03 

efficiency  of  speaker,  13-10 

integral,  5-35 

mks  unit,  1-46 

of  mechanical  and  electrical  systems,  5-57 

symbol,  1-46,  1-73 

volume,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 

Engineering  terms,  abbreviations,  1-71 
English  candles,  conversion  factors,  1-65 

Legal  Standard  Wire  Gage,  1-69 

system  of  units,  1-42 
Envelope  delay,  5-36 

in  f-rn  distortion,  8-26 

double-cavity  resonator,  graphical  symbol,  1-77 

gas-filled,  graphical  symbol,  1-77 

high-vacuum,  1-77 

shield  within,  graphical  symbol,  1-78 

single-cavity     resonator,     graphical     symbol, 

1-78 

Epstein  test,  2-75 
Equalizers,  attenuation,  16-28 

in  passive  circuits,  6-05 

loss,  5-16,  5-18 

phase,  5-16,  5-21 
Equalizing  pulses,  20-13,  20-17 
Equally  tempered  scale,  11-09 
Equation,  abbreviation,  1-71 

algebraic,  1-02 

Bessel's,  1-15 

calculus,  1-12 

circuit,  5-05 

common  curves,  1-05 

containing  complex  quantities,  1-07 

cubic,  1-03 

differential,  1-13 

exponential,  1-10 

hyperbolic,  1-10 

Maxwell,  1-45,  5-50 

mesh,  5-05 

nodal,  5-06 

of  common  curves,  1-05 

quadratic,  1-03 

simultaneous,  1-03 
linear,  1-04 

trigonometric,  1-07 

wave,  5-51 

Equilateral  hyperbola,  equation,  1-05 
Equipment,  telegraph,  see  Telegraph  equipment 
Equipotential  cathode,  4-04 
Equivalent  circuit,  of  crystal  oscillator  circuit, 

7-93 

of  simple  antenna,  6-67 
of  triode,  6-08 

impedance,  6-08 

negative  resistance,  5-49 

network,     complete,     audio-frequency    trans- 
former of,  6-14 

-noise  resistances,  4-21 

primary  impedance,  6-08 

quadripoles,  5-13 

Ergodic  state  of  acoustics  of  rooms,  12-41 
Ergs,  1-44,  1-46 

conversion  factors,  1-57,  1-58 
Errors  of  observation,  1-15 
Erythema,  23-07 
Ethocel,  2-37 
Ethofoil,  2-37 
Ethyl  abietate,  dielectric  properties,  2-49 

acetate,  dielectric  properties,  2-49 

alcohol,  dielectric  properties,  2-49 


INDEX 


17 


Ethyl  alcohol,  relation  between  dielectric  constant 
and  resistivity,  2-51 

benzene,  dielectric  properties,  2-49 

cellulose,  2-37 

power  factor  at  high  frequencies,  2-34 
properties,  2-26 

ether,  dielectric  properties,  2-49 
Ethylene,  dielectric  constant,  2-54 

glycol,  dielectric  properties,  2-49 
Eureka,  properties,  2-05 
Eustachian  tube  of  ear,  12-02 
Evanohm,  properties,  2-05 
Excello,  properties,  2-05 
Excelsior,  properties,  2-05 
Exchange  telephone  plant,  17-78 

cables,  17-93 

design,  17-92 

effective  transmission  performance,  17-79 

Icop  losses,  17-80 

open- wire  facilities,  17-96 

service  requirements,  17-78 
Exciter  lamp,  16-22 
Excitor,  graphical  symbol,  1-77 
Excitrons,  4-80 

defined,  4-75 
Exhaust  fan  noise,  11-64 
Exponential  formulas,  1-10 

function  of  x,  1-10 

horn,  13-05 

tables,  1-26 

wave  form,  repetition  rate  of  pulses  determined 

by,  9-19 

Exposure  meter,  photographic,  15-18 
External,  abbreviation,  1-71 
Extrapolation,  4-09 
Eyes,  aberrations,  14-29 

depth  of  focus,  14-30 

movement,  14-28 

optical  characteristics,  14-29 

refractive  errors,  14-30 

refractive  media,  14-26 

resolving  power,  14-40 

retinal  rivalry,  14-48 

schematic,  14-28 

structure,  14-25 

visual  acuity,  14-40 

F  layer,  10-38 
Facsimile,  18-38,  19-02 
duplicators,  19-23 
picture  elements  in,  19-02 
reception  of,  19-02 
recorder,  phasing,  19-21 
recording  amplifiers,  19-16 
recording  systems,  19-11 

carbon-paper,  19-15 

comparison,  19-16 

dry  electrolytic,  19-13 

photographic,  19-11 

wet  electrolytic,  19-12 
scanner  amplifiers,  19-07 

frequency  spectrum,  19-07 

types,  19-08 

typos  of  signals,  19-08 
scanning  systems,  19-02 

methods  used  in,  19-03 
synchronization,  19-18 

magnetic  brake,  19-19 

start-stop,  19-20 
tape  systems,  19-23 
transmission  of,  19-02 

radio,  19-22 

wire  line,  19-22 
Factories,  noise  levels  acceptable  in,  12-58 


Factors,  conversion,  1-47 
Factory  broadcast  system,  16-16] 
Fading,  10-45 

reduction  of,  7-125,  18-57 
Fahy  penneameter,  2-74 
Farad,  1-44,  1-45,  1-46 

abbreviation,  1-71 

conversion  factors,  1-62 
Faraday  constant,  1-79 

dark  space,  4-59 

Faradays,  conversion  factors,  1-58 
Faraday's  law,  5-50 
Fahrenheit  units,  1-43 

conversion  table,  1-64 
Fathom,  1-47 
Fault  location,  11-41 
FCC  Standards,  television,  20-20 

wave  form,  television,  20-17 
Feedback,  negative,  7-31,  7-51 

of  reverberation,  12-47 
Feeling,  threshold  of,  12-09 
Feet,  see  also  Foot 

conversion  table,  1^7 

cubic,  conversion  table,  1-49,  1-50 

square,  conversion  table,  1-48 
Felt,  hair,  12-63 
Fermat's  principle,  14-02 
Ferroelectric  crystals,  13-57 

defined,  13-57 

Ferromagnetic  materials,  2-57 
Ferro-nickel,  properties,  2-05,  2-10 
Ferrous-cored  inductor,  3-42 
Ferry-Porter  law,  14-35 

Fiber  boards,  coefficients  of  sound  transmission, 
12-65 

bone,  properties,  2-26 

commercial,  properties,  2-26 

vulcanized,  2-37 
Fiberglas,  2-37 
Fibestos,  2-36,  2-37 
Fibron,  2-37 
Fidelity,   determination  of,  in  superheterodyne 

receivers,  7-56 
Field,  20-06 

frequency,  20-20 

intensity,  6-70,  10-31 

of  view,  20-02 

sequential  television,  20-65 
Fields,  due  to  a  current  in  a  wire,  5-52 
Fifth,  11-08 
Figure  of  merit,  3-31 
Filament,  defined,  4-04 

current,  4-08 
denned,  4-05 

power,  7-106 

transformer,  6-26 

voltage,  changes  in,  4-26 

denned,  4-05 

Filamentary  cathode,  4-08 
Filters,  see  also  Wave  filters 

attenuating  band  of,  6-33 

B  supply,  7-108 

band-pass,  6-33,  6-58 

attenuation   (Tchebycheff  type)   character- 
istic, 6-58 
illustrative  design,  6-59 

constant-.^  section,  6-48 

cutoff  frequencies  and,  6-33 

design,  Tchebycheff  type  characteristics,  6-56 

elementary   symmetrical   band-pass  sections, 
6-44,  6-45,  6-46,  6-47 

frequency  control  factors  required  for  specified 
reflection  coefficient,  6-57 

general  composite,  6-40 


18 


INDEX 


Filters,  general  composite,  elementary  constitu- 
ents, 6-49 
high-pass,  6-33,  6-58 

constant-J£  type  image  impedance,  6-50 
idealized,  5-35 

image  impedance  of  general  composite,  6-49 
image  impedance  (Tchebycheff  type)  character- 
istic, 6-58 

image  impedance  theorem,  6-39 
insertion  loas  for  frequencies,  6-33 
low-pass,  6-33,  6-58 

constant-J?  type  image  impedance,  6-50 
m-derived  sections,  6-48,  6-50 
TO-derived  terminating  sections,  6-54 

design  information  for,  6-52 
open-circuit  transfer  impedance,  6-38 
pass  band  of,  6-33 

image  impedance  and,  6-36 
simple  section,   mid-series  constant-^  imped- 
ance, 6-51 

mid-shunt  constant- K  impedance,  6-51 
slope,  8-19 
symmetrical,  6-41 

high-pass  sections,  6-43 
image  transfer  function,  6-56 
lattice  configuration,  6-41 
low-pass  sections,  6-43 
symmetrical  sections,  6-39,  6-41 
elementary  structures,  6-48 
lattice,  6-41 

terminating  sections   having  two-image    con- 
trolling frequencies,  6-54,  6-55 
transfer  constant  theorem,  6-39 
two-frequency  control  image  impedance  of  un- 

syrnmetrical  sections,  6-53 
two-frequency  control  sections,  6-53 
types,  6-33 

unsymmetrical  sections,  6-50 
x-ray,  23-12 
First  aid,  23-17 
Fish  paper,  2-37 

Fisher-Hinnen  method  of  analysis,  5-47 
Fixed  capacitance  condensers,  3-60 
Flamenol,  2-37,  2-48 
Flashover,  2-24 
Flat  field,  20-04 

selectivity  curve,  f-m  distortion  and,  8-28 
-top  antennas,  6-62,  6-80 
-type  resistors,  3-09 
Flax-Ii-num,  compliance  and  resistance  data  in 

insulation  of  vibration,  12-62 
Flexible  coupling  units,  10-18 

cushions  in  sound  insulation,  12-60,  12-61 
panels,  12-62 
resistors,  3-09 
wave  guides,  10-18 
Flexural  strength,  2-28 

effect,  4-20 
Flicker,  20-05,  20-66 

requirements,  20-06 
Flip-flop  circuits,  of  counters  and  computers,  9-13, 

9-14 

stable  in  either  condition,  illustrated,  9-17 
Floor  partitions,  12-66 
Fluctuation  current,  4-20 

noise  interference,  divisional  noise  of  vacuum 

tubes,  8-30 
in  f-m  systems,  8-30 
shot  noise  of  vacuum  tubes,  8-30 
thermal  noise,  8-30 
voltage,  4-21 

Fluid  ounce,  conversion  factors,  1-49,  1-50 
Fluorescence,  denned,  15-30 
Fluorescent  lamps,  15-37 


Fluoroscopy,  tubes,  4-89 
Fluor s,  denned,  15-30 
Flute,  power  of,  12-25 
Flutter  echo,  12-40 

Flutters,  acoustic  properties  of  rooms  and,  12-40 
Flux  density,  electric,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 
magnetic,  cgs  unit,  1-46 
conversion  table,  1-63 
mks  unit,  1-46 
symbol,  1-46,  1-73 
of  power  transformers,  6-28 
dielectric,  conversion  table,  1-58 
displacement  f.,  symbol,  1-73 
electric,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 
-linkage,  symbol,  1-73 
-magnetic,  cgs  unit,  1-46 
conversion  table,  1-63 
denned,  2-60 
mks  unit,  1-46 
symbol,  1-46 
voltmeter,  2-76 

F-m  distortion,  a-m  rejection  and,  8-29 
amplitude  modulation,  8-26 
balanced  discriminator  and,  8-28 
commercial  entertainment  receivers  and,  8-27 
delay  characteristics,  8-26 
downward  amplitude  modulation  and,  8-29 
due   to   incomplete   rejection   of   amplitude 

modulation,  8-27 
due  to  multipath  reception,  8-29 
envelope  delay  in,  8-26 
flat  selectivity  curve  and,  8-28 
f-m  detector  input-output  characteristic,  8-28 
from  amplitude  modulation,  8-28 
from    three    cascade    double-tuned    trans- 
formers, 8-27 
harmonic  distortion,  8-28 
in  f-m  systems,  8-26 
inadequate  bandwidth,  8-26 
incomplete  rejection,  8-26 
input  capacity  of  amplifier  tubes  and,  8-32 
modified   analysis,   method   for   evaluating, 

8-27 

multipath  distortion  effects,  8-29 
multipath  transmission,  8-26 
non-linear  phase  characteristic,  8-26 
non-uniform  delay  and,  8-26 
null  in  transmission  and,  8-29 
quasi-steady-state  approximation,  8-26 
second  harmonic,  8-28 
side  responses,  reduction  of,  8-29 
sideband  analysis-synthesis  method  for  eval- 
uating, 8-27 

skip  transmission  and,  8-29 
small  deviation  ratio,  8-27 
small  distortion  in  systems  with,  8-27 
third  harmonic,  8-28 
equipment,  measurement  of  frequency  swing, 

8-08 
receivers,  8-16 

a-m  receivers  and,  8-16 

commercial    entertainment,    f-m    distortion 

and,  8-27 

frequency  detectors  of,  8-19 
limiters,  grid-bias,  8-24 
multiple  tuning  positions,  8-17 
superheterodyne  principle,  8-16 
systems,  adjacent-channel  interference  in,  8-30 
beatnote  and,  8-30 
co-channel  interference  in,  8-30 


INDEX 


19 


F-m.  systems,   de-emphasis  low-pass  filter  and, 

8-31 

fluctuation  noise  interference,  8-30 
impulse  noise  interference  in,  8-31,  8-32 
carrier  amplitude  and,  8-32 
"click,"  8-32 

grid-bias  limiter  and,  8-32 
"pop,"  8-32 

pulse  of  phase  modulation  and,  8-32 
step  of  phase  modulation  and,  8-32 
transient  impulse  amplitude  and,  8-32 
interference  in,  8-26,  8-29 

audio  noise,  rms  value  of,  and,  8-31 
fluctuation  noise,  8-29 
f-m  ratio  and,  8-31 
impulse  noise,  8-29 
noise  signals,  8-29 
output  noise  spectrum  and,  8-31 
spurious  receiver  responses,  8-29 
large  deviation  ratio  and,  8-31 
peak  noise,  8-31 

signal-to-interference  ratio,  8-30 
signal-to-noise  ratio  importance  and,  8-31 
small  deviation  ratio  and,  8-31 
threshold  for  f-m  improvement,  8-31 
triangular  f-m  beatnote  spectrum  and,  8-30 
transmitters,  8-09 
circuits,  8-15 

for  emergency  communication,  8-09 
for  f-m  broadcasting,  8-09 
sound  transmitters  for  television  broadcast- 
ing, 8-09 
types,  8-09 

Focusing  of  sound,  12-70 
Foot,  see  also  Feet 
abbreviation,  1-71 
-candle,  abbreviation,  1-71 
cubic,  abbreviation,  1-71 
-Lambert,  abbreviation,  1-71 
-pound,  abbreviation,  1-71 
conversion  factors,  1-57,  1-58 
-second  (system),  abbreviation,  1-71 
square,  abbreviation,  1-72 
Force,  cgs  unit,  1-46 
coercive,  defined,  2-59 
conversion  table,  1-55 
counter  electromotive,  abbreviation,  1-71 
electromotive,  abbreviation,  1-71 
measurement,  3-02 
symbol,  1-73 
unit,  1-44 

factor  of  speaker,  13-12 
magnetizing,  2-57 
cgs  unit,  1-46 
conversion  table,  1-64 
defined,  2-60 
mks  unit,  1-46 
symbol,  1-46 

magnetomotive,  abbreviation,  1-71 
cgs  unit,  1-46 
conversion  table,  1-64 
mks  unit,  1-46 
symbol,  1-46,  1-73 
mks  unit,  1-46 

moment  of,  conversion  table,  1-56 
per  unit  area,  conversion  table,  1-56 
standard  of,  1-42 
symbol,  1-46,  1-73 
Foreign  wire  relations,  10-67 
Formation  voltage,  3-69 
Formex,  2-37,  2-47 
Formica,  2-37 
Forming  voltage,  3-69 
Formulas,  algebraic,  1-02 


Formulas,  approximation,  1-16 

calculus,  1-12 

cutoff  voltage,  4-40 

exponential,  1-10 

hyperbolic,  1-10 

mensuration,  1-17 

temperature  conversion,  1-64 

trigonometric,  1-07 
Formvar,  2-37,  2-47 
Forticel,  2-36,  2-37 
Fortisan,  2-37 
Foster's  theorem  on  driving  point  impedances, 

6-35 
Fourier  integral,  5-27,  5-33 

series,  5-27,  5-33 
Fourier's  theorem,  5-02 
Four-terminal  networks,  5-10,  6-33 

driving  point  impedances,  5-10 

equivalent  quadripoles,  5-13 

image  parameters,  5-13 

lattice  or  bridge,  5-13 

restrictions  for  physical  readability,  5-11 

T  and  TT,  5-13 

terminated,  5-11 

transfer  impedance,  5-10 
Fourth,  11-08 
Frame,  20-06 

frequency,  20-20 
Free  charge,  2-23 
Free-running  multivibrator,  9-18 

space  transmission  law  of  antennas,  6-72 
Freedom,  degrees  of,  5-56 
Freezing  point,  abbreviation,  1-71 
French  horn,  power  of,  12-25 
Freon,  2-56 
Frequencies,  band,  analysis,  11-56 

complex,  5-04 

defined,  11-02 

discrete,  distribution,  11-54 

extremely  high,  1-80 

formulas,  11-03 

high,  1-80 

low,  1-80 

measurements,  11-02 

medium,  1-80 

nomenclature,  1-80 

spectrum  of  electrical  phenomena,  11-14 

super  high,  1-80 

ultra  high,  1-80 
measurement,  11-11 

very  high,  1-80 

very  low,  1-80 
Frequency  allocation,  television,  20-20 

analysis  of  exhaust  fan  noise,  11-64 

angular,  symbol,  1-73 

bandwidth,  required  for  a  certain  speed  of  in- 
formation, 9-04,  20-04 
widening  of  square  pulse  by  reduction,  9-04 

cgs  unit,  1-46 

characteristics,    measurements    of,    audio-fre- 
quency transformer,  6-25 
of  input  and  interstage  transformers,  6-20 

comparison,  11-05 

control,  see  also  Frequency  modulation 
Crosby  method,  8-13 
direct  frequency  modulation,  8-13 
electromechanical,  8-14 
pulse  counter,  8-14 

counters,  9-13 

field,  20-20 

flip-flop  circuits,  9-13 

frame,  20-20 

detectors,  discriminator  characteristics,  8-20 
discriminator  circuits,  8-20 


20 


INDEX 


Frequency  detectors,  discriminators  as,  8-19 

of  f-m  receivers,  8-19 

ratio-type,  8-21 

amplitude  compensation,  8-23 
regulation  curves,  8-22 

side-tuned  circuits  as,  8-21 

slope  filter  as,  8-19 

simple,  8-19 
difference  limen,  of  hearing  with  decibels  above 

threshold  (hearing),  12-10 
discrimination,  11-63 
distortion,  5-16 
distortion,  effects  on  sounds,  12-30 

resonance  type,  effect  on  articulation,  12-33, 

12-34 
dividers,  9-13 

counting  type,  9-13 
division  in  pulse  circuit,  9-13 
effect  on  magnetization,  2-70 
meters,  induction,  11-11 

vibrating-reed,  11-11 

Weston,  11-11 
mks  unit,  1-46 

modulation,  7-71,  8-02,  see  also  Frequency  con- 
trol 

Beasel  functions,  8-04 
values,  8-05 

broadcasting,  8-09,  16-31 
amplitude  noise,  8-10 
coverage,  16-32 
distortion,  8-10 
frequency  characteristic,  8-10 
frequency  control  and  modulation,  8-10 
intermediate  frequency,  8-16 
noise  level,  8-10 
transmitter,  8-10 

by  input  capacitance  variation,  8-12 

by  phase-modulation  method,  8-11 

by  reactance  tubes,  8-12 

degree  of,  8-04- 

detector,  a-m  rejection  in,  11-53 

input-output  characteristic,  distortion  and, 
8-28 

deviation  ratio,  8-03 

direct,  8-12 

discriminator  action  of  detectors,  8-03,  8-19 

distortion,  see  F-m  dintortion 

equivalent  sideband,  8-04 

for  emergency  transmitters,  8-15 

frequency  detector  conversion,  8-17 

frequency  swing,  8-04 

from  non-uniform  amplitude  and  phase  char- 
acteristics, 8-2(5 

fundamental  relations,  8-02 

mathematical  equivalent  of  discriminator  ac- 
tion, 8-09 

measurements,  11-50 

methods,  8-10 

modulation  index,  8-03,  8-04,  8-05 

noise  characteristics,  8-31 

pass  band,  transmitting  and  receiving,  and, 
8-02 

peak  angle  swing  and,  8-06 

receivers,  see  F-m  receivers 

sideband  distribution,  8-06 

sidebands  and,  8-02 

single-,  8-02 

studio-transmitter  link  use  of,  8-02 

systems,  8-02 

theory  of  noise  suppression  and,  8-03 

transmitters,  see  F-m  transmitters 

two-frequency,  8-07 
multipliers,  9-13 
range,  audible,  12-30 


Frequency  response  of  output  transformers,  6-17 

rotational,  symbol,  1-73 

sensitivity,  12-10 

shift,  effects  of,  12-35 

spectrum,  5-28 

of  pulse,  9-11,  9-12 

stability  of  oscillation,  7-92 

standard    used    in    frequency    measurement, 
11-05 

swing  of  frequency  modulation,  8-04 

symbol,  1-46,  1-73 
Fricative  consonants  of  speech,  12-20 

fill  in  visible  speech,  12-21 
Frictional  modulation  in  speech,  12-19 
Frolich-Kennelly  relation,  2-69 
Front  feed  for  reflectors  of  antennas,  6-86 

porch,  20-16 

-wall  effect,  15-13 
Frustum,  volume  of,  1-18 
Full  coaxial  cylinder  cavity  resonator,  7-98,  7-101 

-wave  rectifier,  7-107 
Fullerboard,  2-37 
Furfural  resins,  2-37 
Furlong,  conversion  factors,  1-47 
Fuse,  17-05 
Fusion  point,  abbreviation,  1-71 

G.  E.  Mycalex,  2-38 
Gages,  1-66 

sheet  metal,  1-66 

wire,  1-66,  1-68,  1-69,  1-70,  2-12 

comparison,  1-09,  1-70 
Gain,  control,  16-07 

insertion,  11-32 
Gallium,  properties,  2-05 
Gallon,  abbreviation,  1-71 

British  Imperial,  convcrnion  factors,  1-50 

liquid,  conversion  tablo,  1-19 

U.  S.  (dry  measure),  conversion  factors,  1-50 

XT.  S.  (liquid),  convorHion  factors,  1-50 
Galvanized  Sheet  Gage,  1-66 
Galvanometer,  graphical  symbol ,  1-76 
Gamma  (transfer  characteristic),  20-29 

correction,  20-29 
Gang  tuning  condenser,  7-121 
Gap  spacing,  4-53 
Gas  current,  defined,  4-05 

-filled  oscillators,  deionization  time  of  gas  in 

operation,  7-91 
ionization  time  of  gas  in  operation,  7-91 

-filled  tubes,  4-02 

triodo,  4-59 

tube,  defined,  4-03 

volume,  1-79 

x-ray  tubes,  4-81 
Gaseous  conduction,  4-58 

tubes,  4-58 
Gases,  as  dielectrics,  2-53 

conductivity,  2-54 

dielectric  strength,  2-54 

iomzation,  2-54 
Gauss,  1-46 

conversion  factors,  1-63 
Gelva,  2-37 

General  composite  filters,  6-40 
elementary  constituents,  6-49 

filter  network,  6-33 

-purpose  triode,  4-31 
Generalized  displacement,  5-57 

mass,  5-57 

velocity,  5-57 
Generators,  constant  voltage,  4-07 

double-pulse,  9-24 

multiple-pulse,  9-24 


INDEX 


21 


Generators,  of  constant  current,  4-07 
Geometrical  progression,  1-02 

representation  of  complex  number,  1-06 
Geon,  2-37,  2-47,  2-48 
German  candles,  conversion  factors,  1-65 

silver,  properties,  2-05 
Germanium,  properties,  2-05 
Gilbert,  1-46 

conversion  factors,  1-64 
Gill,  liquid,  conversion  factors,  1-49,  1-50 
Gilsonite,  2-35,  2-37 
Giorgi,  1-45 
Glass,  2-37 

-bonded  mica,  2-38 

power  factor  at  high  frequencies,  2-34 
properties,  2-26 
high  silica,  power  factor  at  high  frequencies, 

2-34 

properties,  2-10,  2-26 

Pyrex,  power  factor  at  high  frequencies,  2-34 
textiles,  2-38 
Glide  path,  22-15 

Glottis  of  vocal  cords,  acoustic  wave  form  pro- 
duced at,  12-19,  12-20 
Glow-discharge  tubes,  4-72 
Glowray,  properties,  2-05 
Glycerin,  dielectric  properties,  2-49 
Glyptal,  2-34T  2-3S 
Gold,  properties,  2-05,  2-10 
Goldschmidt  alternators,  7-94 
Goniometer,  22-08 
Gradient,  potential,  cgs  unit,  1-46 
conversion  table,  1-60 
mks  unit,  1-46 
symbol,  1-46 
Grahamite,  2-35 
Grain-oriented  materials,  2-61 
Grains,  conversion  factors,  1-54, 1-55 
Gram,  abbreviation,  1-71 
-calorie,  1-65 

abbreviation,  1-71 
conversion  factors,  1-57,  1-64,  1-65 
-centimeters,  conversion  factors,  1-56 
conversion  factors,  1-54,  1-55 
Graphical  symbols,  1-76 
Graphite  grains,  properties,  2-10 

properties,  2-10 
Gravity,  acceleration,  1-79 

specific,  abbreviation,  1-72 
Greek  alphabet,  use  for  symbols,  1-79 
Grenz  rays,  23-12 
Grid  bias,  denned,  4-05 
-bias  Hmiters,  8-24 
characteristic  curve,  4-10 
control,  defined,  4-04 
current,  defined,  4-05 

detector,  7-79 
defined,  4-04 

driving  power,  defined,  4-05 
emission,  4-11 
defined,  4-05 
graphical  symbol,  1-77 
leak,  7-93 
modulation,  7-73 
-plate  characteristic,  4-11 
-plate  transconductance  (mutual  conductance) , 

5-42 
resistance,  4-11 

maximum  allowable,  4-27 
resistors,  3-11 
screen,  defined,  4-04 
space-charge,  defined,  4-04 
suppressor,  defined,  4-04 
voltage,  defined,  4-05 


Grinders,  10-43 

Gross  ton,  conversion  factors,  1-54 

Ground  controlled  approach,  22-26 

graphical  symbol,  1-76 

return  circuit,  10-87 

system,  antennas  and  use  of,  6-69 

Wagner,  11-26 

wave,  see  Wave  propagation,  ground  wave 

wires,  10-93 
Grounded  capacitors,  11-24 

-grid  amplifiers,  7-31,  7-49 
Group  velocity,  5-36 
Growth  of  sound,  12-42 
Gullstrand  schematic  eye,  14-28 
Gummon,  2-38 

properties,  2-26 

Gunther's  chain,  conversion  factors,  1-47 
Gutta-percha,  2-38 

properties,  2-26 

Hair  felt,  12-53,  12-56 
Half -wave  antenna,  6-71 
Halowax,  2-38 

Hand,  conversion  factors,  1-47 
Hard  plaster,  12-55 
rubber,  2-38 
x-rays,  4-82 
Harmonic    amplifiers,    radio    transmitters    and, 

7-130 
analysis,  of  current  for  a  sinusoidal  applied 

voltage,  5-46 

distortion  in  f-m  systems,  S-28 
Harmonics,   of   the   intermediate  frequency   of 

superheterodyne  receivers,  7-122 
Hartley  oscillators,  7-93 
Heading,  defined,  22-05 
Heaped  bushel,  conversion  factors,  1-50 
Hearing,  average  loss,  12-08 
binaural  vs.  monaural,  12-06 
difference  limen,  12-09 
limits  of,  12-08 
mechanism  of,  12-02 
monaural,  16-02 

of  speech,  effect  of  noise  and,  12-72 
effect  of  reverberation  on,  12-73 
in  auditoriums,  12-69 
sense  of,  12-02 
tests  with,  losses  greater  than  25  db  and  45  db, 

12-09 

variation  with  age,  12-07 
Heart  disease,  diagnosis,  23-09  _ 

human,  relaxation  oscillation  in,  7-84 

sound  of,  23-09 
Heat,  conversion  table,  1-57 
electrical  equivalent,  1-79 
mechanical  equivalent,  1-79 
quantity  of,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 
specific,  abbreviation,  1-72 

conversion  table,  1-64 
Heater  current,  4-08 

denned,  4-05 
defined,  4-04 
delay,  7-114 
graphical  symbol,  1-77 
voltage,  changes  in,  4-26 

denned,  4-05 

Heating  effect  of  r-f  current,  23-04 
of  power  transformer,  6-29 
time,  cathode,  4-08 
Heaviside  layer,  10-37 

Lorentz  system  of  units,  1-43 
operational  calculus,  5-34 


INDEX 


Heaviside,  unit  function,  5-27 
Hectare,  conversion  factors,  1-48 
Hectogram,  conversion  factors,  1-54 
Hectoliter,  conversion  factors,  1-49 
Hectometer,  conversion  factors,  1-47 
Hectometric  waves,  1-80 
Hefners,  conversion  factors,  1-65 
Height,  effective,  6-70 
Heising  modulation,  7-74 
Helicotrema  of  ear,  12-03 
Helium,  minimum  sparking  potentials,  2-54 
Hemispheres,  conversion  table,  1-51 
Hemit,  2-38 

properties,  2-26 
Henry,  1-44,  1-45,  1-46 

abbreviation,  1-71 

conversion  factors,  1-63 
Heptane,  dielectric  properties,  2-49 
Heptode,  denned,  4-04 
Hercules  EC,  2-37 
Herculite,  2-38 

Heterodyne  analyzer,  11-61,  11-66 
Heterophoria,  14-29 
Heusler  alloys,  2-57 
Hexane,  dielectric  properties,  2-49 
Hexode,  defined,  4-04 
High-fidelity  broadcasting,  16-33 

-frequency  antenna,  6-82 

-frequency  broadcasting,  16-31 

compensation  methods  of  wide-band  ampli- 
fiers, 7-43 
surgery,  23-06 
apparatus,  23-04 

-H  per mea meter,  2-74 

-inductance  antenna  coupling  of  radio  receivers, 
7-115 

level  modulation,  7-74 

-pass  filter,  6-33 

-permeability  materials,  2-60 

-power  audio  amplifiers,  7-22 

-power  radio  transmitters,  7-136 
audio  equipment,  7-136 

~Q  circuits  of  i-f  amplifiers,  7-62 

resistances,  3-17 

transmission,  10-05 

-vacuum  tubes,  4-02 
defined,  4-04 

-voltage  shock,  23-17 

-voltage  therapy  tubes,  4-85 
Highlight  brightness,  20-03 
Hinge  joints  of  wave  guides,  10-26 
Hipernik,  properties,  2-06 
Hiss  tone  in  speech,  12-21 
Hole-and-slot  magnetrons,  4-42 
Homes,  noise  levels  acceptable  in,  12-58 
Hopkinson  alloy,  properties,  2-06 
Horizontal  hyperbola,  1-05 

parabola,  1-05 

polarization,  20-20 

resolution,  20-04 

synchronizing  pulse,  20-17 
Horizontally  polarized  antenna,  6-85 
Horns,  6-79,  13-05,  13-10 

biconical,  6-80 

catenoidal,  13-06 

conical,  6-80 

defined,  13-05 

directional  properties,  13-08 

electromagnetic,  6-79 

forms  of,  6-80 

pyramidal,  6-80 

quasi-optical,  directivity  of,  6-76 

Salmon,  13-06 

sectoral,  6-80 


Horsepower,  abbreviation,  1-71 

conversion  factors,  1-58 

-hour,  abbreviation,  1-71 
conversion  factors,  1-57 

metric,  conversion  factors,  1-58 
Hospitals,  noise  levels  acceptable  in,  12-58 
Hot-cathode  thyratrons,  4-71 

x-ray  tubes,  4-81 
Hotels,  noise  levels  acceptable  in,  12-58 

sound  insulation,  12-57 
Hour,  abbreviation,  1-71 

conversion  table,  1-51 

Humidity,  variations  in,  use  of  stable  fixed  con- 
densers and  tuning  by  iron-core  coils,  7-60 
Hundred,  abbreviation,  1-71 
Hundredweight,  conversion  factors,  1-54 
Huygens'  principle  of  wave  propagation,  6-76 
Hybrid  coils,  see  Transformers,  three  winding 

junctions,  11-75 
Hycar,  2-38 

OR,  2-35 
Hydrogen,  acoustic  properties,  13-02 

dielectric  constant,  2-54 

dielectric  properties,  2-49 

minimum  sparking  potentials,  2-54 
Hyperbola,  equation,  1-05 
Hyperbolic  cosine,  abbreviation,  1-71 

formulas,  1-10 

functions,  1-10 
chart,  1-26 

relations   between   trigonometric   functions 
and,  1-11 

logarithms,  1-19 

radians,  1-26 

sine,  abbreviation,  1-71 

tables,  1-26 

tangent,  abbreviation,  1-71 
Hysteresis  loops,  2-58 
Hytemco,  properties,  2-06 

la  la,  properties,  2-06 
Ice  load,  10-51 

point,  1-79 
Iconoscope,  15-21 
construction,  15-21 
operation,  15-22 
resolution,  15-24 
sensitivity,  15-24 
signal-to-noise  ratio,  15-24 
studio  television  camera,  20-23 
uniformity,  15-25 
Ideal,  properties,  2-06 

transducer,  5-15 
Idealized  filters,  5-35 
I-f  amplifiers,  7-56 

as  source  of  gain  and  selectivity  in  radio  re- 
ceiver, 7-58 
broad,  7-60 

coefficients  of  coupling,  7-59 
critically  coupled  circuits,  7-59 
double-tuned  stage,  7-63 
flat-topped  selectivity  curve,  7-59 
for  a-m  broadcast  receivers,  7-58 
for  f-m  receivers,  7-61 
medium  bandwidth,  7-58 
narrow  bandwidth,  7-58 
of  superheterodyne  receiver,  7-121 
opposing  couplings,  gain,  and  bandwidth  of 

i-f  stage,  7-60 
opposing  inductive  and  capacitive  coupling 

and,  7-60 

pulse  technique  of,  illustrated,  9-14 
r-f,  7-129 
tuning  stability,  7-60 


INDEX 


23 


I-f  amplifiers,  variable  selectivity,  7-59 

wide-band,  7-63 
gain  and  bandwidth,  11-53 
harmonics,  feedback  of,  7-61 
sensitivity,  defined,  11-50 
Igniter,  defined,  4-75 

graphical  symbol,  1-77 
Ignitrons,  4-78 
contactors,  21-11 
defined,  4-75 

ignitor  characteristics,  4-80 
rectifiers,  21-10 
tests,  4-80 

Ilium,  properties,  2-06 
Illuminance,  14-16 
Illumination,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 
visual  acuity  and,  14-42 
Image,  characteristics,  15-44 
dissector,  15-19 
resolution,  15-21 
sensitivity  of,  15-20 
signal-to-noise  ratio,  15-21 
uniformity  of,  15-21 
impedances,  5-13,  6-37 

characteristics  of  m^-derived  filter  sections, 

6-53 

of  general  composite  filter,  6-49 
of  two-frequency  control  sections  of  filters, 

6-53 

theorem  of  filters,  6-39 
wave-propagation  theory  and,  6-33 
orthicon,  15-27 
resolution,  15-29 
sensitivity,  15-29 
studio  television  camera,  20-23 
uniformity,  15-29 
parameters,  5-13 

coincidence  conditions,  6-36 
properties,  6-36 
summary  of  properties,  6-38 
response  of  superheterodyne  receivers,  7-122 
transfer  constant,  6-33 

wave-propagation  theory  and,  6-33 
transfer  functions,  of  networks,  6-37 

of  symmetrical  filter,  6-56 
Imaginary  period,  1-10 
quantity,  1-06 
unit,  1-06 

Impact  sounds  of  noise  conduction,  12-60 
Impedance,  blocked,  5-65 
bridge,  in  circuits,  6-12 
characteristic,  10-02 

of  uniform  line,  5-25 
circle,  5-65 

common,  in  coupled  circuits,  6-06 
complementary,  5-10 
damped,  5-65 
driving  point,  5-06,  5-10 
imago,  5-B ;  see  also  Image  impedances 
input,  of  uniform  line,  5-26 
inverse,  5-09 
irregularities,  11-39 
lino,  a-c  bridge  method  of  locating  irregularities 

in,  11-39 

matched,  in  single-mesh  circuits,  6-06 
mutual,  5-06 

non-matched,  in  single-mesh  circuits,  6-06 
loads,  5-43 

mechanical,  5-59,  6-03 
motional,  5-66 

mutual,  in  coupled  circuits,  6-06,  6-07 
symbol,  1-73 


Impedance,  normal,  5-65 
open-circuit  transfer,  6-38 
reciprocal,  5-09 

second  image,  method  of  obtaining,  6-40 
self  symbol,  1-73 
short-  and  open-circuit,  6-37 
symbol,  1-73 
transfer,  5-06,  5-10 

transformer  for  wave-guide  component,  10-22 
vector,  5-59 
two-terminal,  5-07 
Imperial  Standard  Wire  Gage,  1-66,  1-69 

units,  1-42,  1-50 
Impulse  noise  interference  in  f-m  systems,  8-31, 

8-32 

Inadequate  coupling,  6-09 
Inch,  abbreviation,  1-71 

circular,  conversion  factors,  1-48 
conversion  table,  1-47 
cubic,  abbreviation,  1-71 

conversion  table,  1-49,  1-50 
miner's,  conversion  table,  1-52 
-pound,  abbreviation,  1-71 
square,  abbreviation,  1-72 

conversion  table,  1-48 
Incidence,  plane  of,  5-53 
Incident  component,  5-23 

waves,  symmetrical  networks,  5-23 

uniform  lines,  5-25 
Inconel,  properties,  2-06 
Incremental  permeability,  defined,  2-60 
Index  of  refraction,  5-53 
Indirectly  heated  cathode,  4-03 
Indium,  properties,  2-06 
Indoor  address  systems,  16-15 
Inductance,  coupling,  6-07 
leakage,  6-14,  6-23 
magnetic,  cgs  unit,  1-46 
conversion  table,  1-63 
mka  unit,  1-46 
symbol,  1-46,  1-73 
measurement,  11-27 

audio-frequency  transformers,  6-25 
mutual,  1-73,  6-07,  6-13 
self,  1-73 
standards,  11-22 
transmission  line,  10-03 
variation,  5-40 
Induction,  intrinsic,  2-57 

defined,  2-59 
frequency  meter,  11-11 
low-frequency,  10-88 

acoustic  disturbance  produced  by,  10-92 
control,  10-93 
coupling  factors,  10-89 
magnetic,  2-57 
defined,  2-59 
symbol,  1-73 
normal,  denned,  2-60 
residual,  defined,  2-60 
saturation,  defined,  2-60 

Inductive  coordination,  for  supply  and  communi- 
cation companies,  10-73 
coupling,  10-78,  10-89 
influence,  10-75 
susceptiveness,  10-80 
winding,  11-19 

Inductometers,  denned,  11-23 
Inductor  iron  core,  graphical  symbol,  1-76 
Inductors,  air-core,  3-31 

properties,  3-31 
best  coH  shape,  3-33 
coil  losses,  3-32 
design  formulas,  3-38 


24 


INDEX 


Inductors,  distributed  capacitance,  3-32 
electrical  design  considerations,  3-32 
ferrous-cored,  3-42 
figure  of  merit,  3-31 
form  materials,  3-36 
graphical  symbol,  1-76 
high-frequency  powdered-core,  3-50 
impregnation,  3-37 
low-frequency  sheet-core,  3-42 
mechanical  design  considerations,  3-36 
power  factor,  3-31 
progressive  universal  winding,  3-34 
shielding,  3-35 
solenoid  windings,  3-33 
specification,  3-38 
time  constant,  3-31 
universal  winding,  3-34 
wire  insulation,  types,  3-32 
Industrial  buildings,  sound  insulation  in.  12-57 

roentgonography,  tubes  for,  4-89 
Inerteen,  2-35 

Inertia,  moment  of,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 

Inflection  in  speech  sounds,  12-20 
Influence,  inductive,  10-75 
Information,  speed  of,  5-28 
Infra-black  level,  20-16 
Infrared  radiation,  medical  uses,  23-06 

therapeutic  use,  23-08 
Initial  permeability,  defined,  2-60 
Inner  ear,  12-03 
In-phase  amplifiers,  7-31,  7-50 
Input  admittance,  conductive  component,  4-16 
compensating  for  changes,  4-19 
of  vacuum  tubes,  4-15 
reactive  component,  4-1 S 
capacitance,  4-19 
defined,  4-06 
variation,  4-19 

frequency  modulation  by,  8-12 
conductance,  4-16,  4-22 
gap,  4-51 
impedance,  of  a  triode,  5-49 

of  uniform  line,  5-26 
reaistance,  7-17 
transformers,  6-19 
leakage  resonance,  6-21 
made  with  two-logged  core,  6-22 
pick-up  of,  6-21 
shielding  of,  6-21 
turns  ratio  of,  6-20 
use  of  shielding  cans  and,  6-21 
Insertion  loss  factor,  5-15 
loss  for  frequencies  in  filters,  6-33 
phase  measurement,  11-34 
phase  shift,  5-16 
Instantaneous  values,  1-74 
Instrument  landing  system,  22-14 
Insulating  liquids,  synthetic,  2-52 
materials,  2-02,  2-21 

dielectric  properties,  2-21 
gases  as  dielectrics,  2-53 
liquid  dielectrics,  2-48 
oils,  2-50 
phenolic,  2-41 
solid  dielectric,  2-25 
test  methods,  2-24 
Insulation,   calculation   of,   in   building   design, 

12-69 

resistance,  2-23 
sound,  see  Sound  insulation 
value  of  rigid  materials,  12-64 


Insulite,  compliance  and  resistance  data  in  insu- 
lation of  vibration,  12-62 
sound  transmission,  12-65 
Insurok,  2-38 
Intake  transformers,  use  of  shielding  cans  and, 

6-21 

Integral  operator,  5-29 
Integrals,  1-12 
Integration,  by  parts,  1-12 

constants,  1-13 
Intelligibility  test,  12-28 
Intensifying  screen,  23-15 
Intensity  difference  limon,  with  decibels  above 

threshold  (hearing),  12-10 
electric,  symbol,  1-73 
electric  field,  cgs  unit,  1-46 
conversion  table,  1-60 
mks  unit,  1-46 
symbol,  1-46 
level  of  sound,  16-13 
luminous,  1-42,  14-15 
cgs  unit,  1-46 
mks  unit,  1-46 
standard  of,  1-43 
symbol,  1-46 
magnetic,  symbol,  1-73 
magnetic  field,  cgs  unit,  1-46 
conversion  table,  1-64 
mks  unit,  1-46 
symbol,  1-46 
sound,  12-04,  12-57 
Interaction  factor,  5-15 

of  electromagnetic  systems,  5-65 
loss,  5-16 

Intercept  delay,  5-36 
Interelectrode  capacitance,  4-14 

defined,  4-06 

Interference,  atmospheric,  10-42 
Interlace,  odd  lino,  20-0(5 
Interlaced  scanning,  20-05 
Intermediate  amplifier,  7-56,  7-129 
frequencies,  "birdiea"  in,  7-57 
choice,  7-56 
examples,  7-57 

harmonics  and  reduction  of  "tweeta"  in,  7-57 
high,  advantage  of,  7-57 
interaction   between    local    oscillation    and 

antenna  circuit  in,  7-57 
subharmonicH,  7-57 
frequency  amplifier,  7-56 
Internal,  abbreviation,  1-71 
International  angntrom  unit,  1-79 
candle,  1-43 

conversion  factors,  1-65 
joule,  1-57 

units  of  electrical  measure,  1-44 
Interpolation  methods,  used  in  frequency  meas- 
urement, 11-10 
Interstage  transformers,  6-19 
leakage  resonance  of,  6-21 
pick-up  of,  6-21 
shielding  of,  6-21 
turns  ratio  of,  6-20 
Intrinsic  induction,  2-57 

defined,  2-59 
Invar,  properties,  2-06 
Invariant,  properties,  2-06 
Inverse  anode  voltage,  4-05 
distance  field,  10-30 
hyperbolic  sine,  1-10 
impedances,  5-09 
sine,  1-08 

Inverted-V  antennas,  6-83 
Inverters,  controlled,  21-09,  21-10 


INDEX 


25 


lonization  current,  4-10,  4-11 

of  gases,  2-54 

time  of  gas-filled  oscillators,  7-91 
Ionosphere,  10-37 
lonthophoresis,  23-03 
Indium,  properties,  2-06 
Iron,  2-61 

-aluminum  alloys,  2-62 

-cobalt  alloys,  2-62 

-core  coils,  use  in  i-f  amplifiers,  7-58 

-core-transformer,  6-13 

effect  of  heat  treatment  on  magnetic  properties, 
2-61 

gage,  1-67 

loss,  2-59 

-molybdenum,  2-62 

-nickel  alloys,  2-62 

oxide,  properties,  2-10 

properties,  2-06,  2-10 

-silicon  alloys,  2-61 
Irradiation,  14-42 
Isolantite,  2-38 

properties,  2-26 

Iso-masking  intensity  of  sound,  12-11 
Isotropic  dielectrics,  5-53 

insulating  medium,  5-51 
IT  calorie,  1-57 
Ivory,  properties,  2-26 

Jack,  graphical  symbol,  1-76 
Joint  use  of  poles,  10-69 
Joule,  1-44 

abbreviation,  1-71 

conversion  factors,  1-55,  1-57,  1-64,  1-65 
Jute,  2-38 

Kanthal,  properties,  2-06 

Kaolin,  2-38 

Karma,  properties,  2-06 

Kataphoresis,  23-03 

Kennelly-Heaviside  layer,  10-37 

Kerosene,  dielectric  properties,  2-49 

Key,  graphical  symbol,  1-76 

Keystone  correction  in  scanning,  20-11 

Kg  cal,  1-46 

Kilocycles  per  second,  abbreviation,  1-71 

Kilogram,  abbreviation,  1-71 

-calorie,  abbreviation,  1-71 

conversion  factors,  1-57,  1-58,  1-64,  1-65 

conversion  factors,  1-54,  1-55 

-meter,  abbreviation,  1-71 
conversion  factors,  1-56 

standard,  1-42 

Kilolines,  conversion  factors,  1-63 
Kiloliter,  abbreviation,  1-71 

conversion  factors,  1-49 
Kilometer,  abbreviation,  1-71 

conversion  table,  1-47 

square,  abbreviation,  1-72 

conversion  table,  1-48 
Kilometric  waves,  1-80 
Kilovolt,  abbreviation,  1-71 

-ampere,  abbreviation,  1-71 
reactive,  abbreviation,  1-72 

conversion  factors,  1-60 
Kilowatt,  abbreviation,  1-71 

conversion  factors,  1-58 
Kilowatt-hour,  abbreviation,  1-71 

conversion  factors,  1-57,  1-64,  1-65 
Kinescopes,  defined,  15-46 
Kinetic  energy,  5-57 
Klystron  amplifier,  4-51 

frequency  multiplier,  4-52 
Klystrons,  4-51 


Klystrons,  defined,  4-51 

employing  transit  time  bunching,  4-51 

integral  cavity  type,  4-51 

reflex,  4-54 

Knife,  high  frequency,  23-05 
Knots,  conversion  table,  1-47,  1-52 
Koroseal,  2-48 
Kriston,  2-34,  2-38 
Kromax,  properties,  2-06 
Kromore,  properties,  2-06 
Krupp  metal,  properties,  2-06 
Kryptol,  properties,  2-10 

L  to  C  ratio1,  of  a  tuned  circuit,  7-87 

of  oscillatory  circuit,  7-88 
Lagrange's  principle,  5-59 
Lambert,  abbreviation,  1-71 
Lambert's  law,  14-05 
Lamicoid,  2-38 
Laminae,  2-39 
Laminates,  2-38 

Lamp  in  pulse  measurements,  9-12 
Lamps,  ballast,  3-22 

exciter,  16-22 

fluorescent,  15-37 

resistance,  3-22  * 

Lanac,  22-28  / 

Land  measure,  area,  conversion  table,  1^8 

length,  conversion  table,  1-47  r 

Laplace  transform,  5-34 
Larynx,  12-19 

artificial,  12-21 

speech  and,  12-19 
Latex,  2-39 

Latitude,  abbreviation,  1-71 
Lattice  networks,  5-13 
Lava,  2-39 

properties,  2-26 
Lavite,  2-39 

properties,  2-26 
Lead-bismuth,  properties,  2-06 

chloride,  properties,  2-10 

-in,  of  radio  antennas,  6-62 

properties,  2-06,  2-10 

-tin  alloy,  properties,  2-10  * 

League  (Great  Britain),  conversion  factors,  1-47 
League  CD".  S-),  conversion  factors,  1-47 
Leakage  current,  4-10 
defined,  4-05 

inductance,  6-23 

measurements    of,    audio-frequency    trans- 
formers, 6-25 
of  audio-frequency  transformer,  6-23 

reactance,  in  driver  transformer,  6-22 

resonance,  of  input  and  interstage  transformers, 

6-21 

Leap  year,  conversion  table,  1-51 
Lecher  wires,  11-13 
Lecture  rooms,  acoustic  design,  12-42 
Legal  units  of  measure,  1-42 
Length,  1-42,  1-46 

cgs  unit,  1-46 

conversion  table,  1-47 

mka  unit,  1-46 

standard  of,  1-42 

symbol,  1-46 
Lenoxite,  2-39 
Lenses,  14-09 

aberrations,  14-11 

bipotential,  14-56 

cathode,  14-58 

compound,  14-10 

cylindrical,  14-54 

dielectric,  use  in  antennas,  6-78 


26 


INDEX 


Lenses,  electrostatic,  14-51 

in  air,  defined,  14-09 

magnetostatic,  14-59 

of  antennas,  6-78 

spherical,  14-54 

thick,  14-54 

thin,  14-10,  14-54 

unipotential,  14-54 
Lepidolite,  2-39 
Lepidomelane,  2-39 
Letter  symbols  for  magnitudes  of  electrical  quan- 

tities, 1-72 
Level,  black,  20-20 

television,  20-13,  20-21 

volume,  1-41 

Libraries,  noise  levels  acceptable  in,  12-58 
Light,  conversion  table,  1-65 

measurement,  14-17 

sense,  14-30 

units,  1-43 

velocity,  1-79 

symbol,  1-73 
...Lighthouse  tube,  7-89 
jcghtning  arrester,  graphical  symbol,  1-76 


.- 


2-39 


,  2-26 


Line,  i 


c       ! 
sequential  ft 

straight,  eau 
synchronizing 


,  10-03 
3 

;       20-65 
' 


amplifier,  7-22  » 
circuits,  5-37  . 

conductor  antem     ' 
detection,  7-79  <• 


,  1-71 

n6tworks'  5'02 
>lex  frequency  plane,  5-04 

ctive,  5-16 
stortion,  5-16 

driving  point  impedances,  5-06 

Duality,  principle  of,  5-07 

Fourier's  theorem,  5-02 

mesh  equations,  5-05 

nodal  equations,  5-06 

non-sinusoidal  currents  and  voltages,  5-02 

power  transfer,  5-15 

single-mesh  circuit,  see  Circuits,  single-mesh 

superposition,  principle  of,  5-02 

theorems,  5-12 

transfer  impedances,  5-06 

two-terminal  impedances,  5-07 
phase,  5-36    . 

simultaneous  equations,  1-04 
velocity,  conversion  table,  1-52 
Xines,  distortionless,  5-26 

program,  radio  telephone  broadcasting,  16-27 
testing,  11-38 

transmission,  see  Transmission  lines 
uniform,  5-24 

characteristic  impedance,  5-25 

incident  and  reflected  waves,  5-25 

input  impedance,  5-26 

propagation  constant,  5-25 

.standing  waves,  5-25 


Lines,  uniform,  voltage  and  current  relations,  5-24 

wire  transmission,  see  Wire  transmission  linen 
Links,  conversion  factors,  1-47 
Linoleum,  12-54 
Linseed  oil,  relation  between  dielectric  constant 

and  resistivity,  2-51 
Lip  mic,  22-12 
Liquid,  abbreviation,  1-71 

dielectric  absorption,  2-51 

dielectric  constant,  2-48 

dielectric  properties,  2-49 

dielectric  strength,  2-51 

dielectrics,  2-48 

gallons,  conversion  table,  1-49 

gill,  conversion  factors,  1-49,  1-50 

measure,  conversion  factors,  1-50 

pints,  conversion  table,  1-49 

quarts,  conversion  table,  1-49 

synthetic  insulating,  2-52 
Lissajous  figures,  11-08 
Liter,  abbreviation,  1-71 

conversion  table,  1-49 
Lithium,  properties,  2-06 
Litz  wire,  3-33 

use  in  i-f  amplifiers,  7-58 
Live  rooms,  12-43 
Loaded  cable  circuits,  10-03 
Loading  map,  10-51 
Loalin,  2-39 
Lobing  method  of  microwaves  in  direction  finding, 

6-87,  6-88 

Localization  of  sound,  12-18,  16-02 
Localizer,  runway,  22-14 
Location  of  impedance  irregularities,  11-39 

of  loudspeakers,  16-14 

of  microphones,  16-05 

of  radio  transmitter,  16-29 
Logarithm  (common),  abbreviation,  1-71 
Logarithm  (natural),  abbreviation,  1-71J 
Logarithmic,  plate  form,  3-57 

voltage  ratio,  1-38 
Logarithms,  1-19 
Lohm,  properties,  2-06 
Lomu,  2-64 
London  Gage,  1-69 
Long  ton,  conversion  factors,  1-54 

waves,  10-33 

-wire  antennas,  6-64 
Longitude,  abbreviation,  1-71 
Longitudinal  circuit  induction,  10-78 
Loop  antennas,  6-62,  6-85 

Bellini-Tosi,    method   of   direction   finding, 

6-88 
sensitivity  to  antenna  effect,  6-87 

coupling,  graphical  symbol,  1-77 

current,  5-26 

hysteresis,  2-58 

output  coupling,  4-47 

voltage,  5-26 
Loran,  22-31,  22-48 

low-frequency,  22-51 

performance,  22-51 

receiving  equipment,  22-50 

sky-wave  synchronized,  22-51 

standard,  22-49 
Loss  angle,  2-22 

equalizers,  5-16,  5-18 

factor,  2-22 

hearing,  with  age,  12-08 

insertion,  5-15 

-phase  relation,  of  a  four-terminal  network,  5-17 

transducer,  5-15 

transition,  5-15 

transmission,  11-33 


INDEX 


27 


Loudness,  computation  for  sounds  with  continu- 
ous energy  spectrums,  12-13 
defined,  12-11 
levels,  12-11 

phon  of,  12-11 
scale,  12-12 
speech,  effect  of,  12-69 

Loudspeaker  loads,  output  transformers,  6-18 
Loudspeakers,  16-13 
defined,  13-08 
efficiency,  13-10 
graphical  symbol,  1-76 
location  of,  16-14 
performance,  13-18 
phasing,  16-14 
tests,  13-18 

two-way  system,  16-23 
types,  16-13 

Low  frequencies,  antennas,  6-80 
Low-frequency  characteristics  of  audio-frequency 

transformers,  6-15 
coordination,  10-88 
induction,  10-88 

acoustic  disturbance  produced  by,  10-92 
control  of,  10-93 
coupling  factors  in,  10-89 
omnidirectional  range,  22-31 
response  of  wide-band  amplifiers,  7-44 
Low-pass  filter,  6-33 
Low-plate-resistance  tube,  7-11 
Low-power  audio  amplifiers,  7-15 
radio  transmitters,  7-136 
resistors,  3-07 

Lower  limit  of  hearing,  12-08 
L-type  networks,  5-13 
Lucero,  properties,  2-06 
Lucite,  2-39,  2-47 
Lumarith,  2-36,  2-37,  2-39 
Lumen,  abbreviation,  1-71 
-hour,  abbreviation,  1-71 
Luminance,  14-16 
Luminescence*  defined,  15-29 
Luminescent  materials,  15-29 
screens,  discharge,  15-44 
ion  spot  in,  15-44 
limitations  imposed  by,  15-43 
size  and  brightness,  15-43 
Luminous  emittance,  14-17 
flux,  ogs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 
intensity,  1-42,  14-15 
cgs  unit,  1-46 
mks  unit,  1-46 
standard  of,  1-43 
symbol,  1-46 
Lungs,  speech  and,  12-19 
Lustron,  2-39 

Machine  noise,  11-63 
M.A.F.,  12-05 
Magnesium  oxide,  2-39 

properties,  2-10,  2-26 
properties,  2-06 
titanate,  properties,  2-30 
Magnetic  armature  speaker,  13-15 
characteristics,  2-57 

measurement,  2-72 
coupling,  5-64 
currents  of  antennas,  6-64 
field  intensity,  cgs  unit,  1-46 
inks  unit,  1-46 
symbol,  1^6 
flux,  ogs  unit,  1-46 


Magnetic  flux,  conversion  table,  1-63 

defined,  2-60 

mks  unit,  1-46 

symbol,  1-46,  1-73 
induction,  2-57 

defined,  2-59 

symbol,  1-73 
intensity,  symbol,  1-73 
materials,  2-57 

effect  of  frequency,  2-70 

effect  of  temperature,  2-69 

high-permeability,  2-60 

magnetic  characteristics,  2-57 
measurement,  2-72 

magnetostriation,  2-70 

permanent  magnet  materials,  2-65 

stress,  2-70 
moment,  cgs  unit,  1-46 

mks  unit,  1-46 

symbol,  1-46,  1-73 
permeability,  1-43 
polarization,  symbol,  1-73 
potential,  conversion  table,  1-64 

symbol,  1-73 
recordings,  13-28 

erasing,  13-28,  13-29 

magnetic  materials  for,  13-36 

recording,  13-28,  13-29,  13-30 

recording  media,  13-35 

reproduction,  13-28,  13-29, 13-33 

scanning,  20-10 
speaker,  13-11 
storms,  10-46 

Magnetization,  cgs  unit,  1-46 
curve,  2-57,  2-68 
mks  unit,  1-46 
symbol,  1-46 
Magnetizing  force,  2-57 
cgs  unit,  1-46 
conversion  table,  1-64 
defined,  2-60 
mks  unit,  1-46 
symbol,  1-46,  1-73 
Magneto  switchboards,  17-03 
Magnetomotive  force,  abbreviation,  1-71 
cgs  unit,  1-46 
conversion  table,  1-64 
mks  unit,  1-46 
symbol,  1-46,  1-73 
Magnetostatic  lenses,  14-59 
Magnetostriction,  2-70 
defined,  7-92 
oscillators,  7-92 
Magnetrons,  4-40 
anode  strapping  methods,  4-46 
cyclotron  frequency,  4-41 
frequency  stability,  4-49 
hole-and-slot,  4-42 
input  characteristics,  1-11 
mode  separation,  4-45 

methods,  4-47 
modes,  r-f  patterns,  4-42 
moding,  4-45 
negative  resistance,  4-40 
non-oscillating,  4-40 
oscillating,  4-40 
output  coupling,  4-47 
scaling,  4-45 
solid  anode,  4-40 
tangential  resonance,  4-42  fn. 
traveling  wave,  4-41 

operation,  4-42 
tuning,  4-50 
Magnets,  permanent,  materials,  2-65 


28 


INDEX 


Magno,  properties,  2-06 
Mahogany,  12-65 
Major  diatonic  scale,  11-09 
Makalot,  2-39 
Male  voices,  11-60 
Malleus  of  eardrum,  12-02 
Manganese-copper,  properties,  2-06 
-nickel,  properties,  2-06 
properties,  2-06 
Manganin,  properties,  2-06 
Mantissa,  of  logarithm,  1-19 
M.A.P.,  12-05 

Marble,  properties,  2-26,  12-55 
Marbon  B,  2-43 

Marine  navigation,  electronic  aids,  22-33 
radio  aids,  22-33 
Consol,  22-52 
Decca,  22-54 

direction-finding  system,  22-39,  22-53 
facsimile,  22-56 
Gee  system,  22-52 
Lanac,  22-53 
Loran,  see  Loran 
miscellaneous  systems,  22-56 
optimum  transmission  parameters,  22-57 
Popi,  22-55 
radar,  see  Radar 

radiobeacon  system,  22-36,  22-38 
Redar,  22-56 
Shoran,  22-52 
Sofar,  22-56 
Sonar,  22-53 
Sonne,  22-52 
Teleran,  22-55 
Marker,  defined,  22-05 

used  in  instrument  landing,  22-17 
Marsh's  patent,  properties,  2-07 
Masking  of  sound,  12-11 
Masonite,  2-39,  12-52 

compliance  and  resistance  data  in  insulation  of 

vibration,  12-62 
Masonite  die  stock,  2-39 

properties,  2-26 
Masonry,  12-55,  12-65 
Mass,  1-42,  1-46 
cgs  unit,  1-46 
conversion  table,  1-54 
electron,  4-14 
mks  unit,  1-46 

per  unit  volume,  conversion  table,  1-55 
resistivity,  1-61 
standard  of,  1-42 
symbol,  1-46 
Master  oscillator,  7-88 

volume  control,  16-07 
Matched   impedances,    in   single-mesh    circuits, 

6-06 

Materials,  cathode,  4-03 
conducting,  2-02 
definitions,  2-02 
specific,  properties,  2-03 
wire  tables,  2-12 

core,  for  pulse  applications,  9-28 
diamagnetic,  2-57 
dielectric,  solid,  2-25 
ferromagnetic,  2-57 
flexible,    coefficients    of    sound    transmission, 

12-65 
compliance  and  resistance  data  in  insulation, 

12-62 

.    grain-oriented,  2-61 
high-permeability,  2-60 
insulating,  2-02 

dielectric  properties,  2-21 


Materials,  insulating,  gases  as  dielectrics,  2-53 
liquid  dielectrics,  2-48 
phenolic,  2-41 
solid  dielectric,  2-25 
test  methods,  2-24 
luminescent,  15-29 
magnetic,  2-57 

effect  of  frequency,  2-70 
effect  of  temperature,  2-69 
high-permeability  materials,  2-60 
magnetic  characteristics,  2-57 

measurement,  2-72 
magnetostriation,  2-70 
permanent-magnet,  2-65 
stress,  2-70 
non-magnetic,  2-64 
non-polar,  2-22 
paramagnetic,  2-57 
permanent-magnet,  2-65 
typical  properties,  2-67 
polar,  2-22 

porous,  sound  insulation  by,  12-62 
properties,  Section  2 
sound-absorptive,  coefficients,  12-48 

practical  considerations,  12-57 
tenebrescent,  15-29 
thin  rigid,  coefficients  of  sound  transmit!" 

12-65 

Mathematical  tables,  1-19 
Mathematics,  1-02 
Maxima,  1-13 

Maximum,  abbreviation,  1-71 
operating  temperature,  2-26 
values,  1-74 
Maxwell,  1-46 

bridge,  2-75,  11-29 
conversion  factors,  1-63 
equations,  1-45,  5-50 

Mayer's  theorem  on  reactive  networks,  6-61 
McBerty  automatic  telephone  system,  17-33 
m-derived  sections  of  filters,  6-48,  6-50 
image  impedance  characteristics,  6-53 
Mean  calorie,  1-57 

square  root,  abbreviation,  1-72 
Measure,  apothecaries'  fluid  conversion  facto. 

1-50 

architect's  conversion  factor,  1-48 
board,  conversion  factor,  1-49 
cubic,  conversion  factor,  1-49 
dry,  conversion  factor,  1-49 
land,  area,  conversion  table,  1-48 
length,  conversion  table,  1-47 
liquid,  conversion  factors,  1-50 
miscellaneous,  conversion  factors,  1-47 
nautical,  conversion  table,  1-47 
ropes  and  cables,  conversion  factors,  1-47 
shipping,  conversion  factor,  1-50 
Measurement,  absorption,  12-58 
capacitance,  11-24 
conductance,  4-12 
current,  11-16 
distance,  9-10 
effective  resistance,  11-27 
electrode  capacitance,  1-14 
inductance,  11-27 
light,  14-17 
mu  factor,  4-13 

of  audio-frequency  transformers,  6-25 
of  magnetic  characteristics,  2-72 
of  noise,  12-57 

primary  electrical  quantities,  11-16 
pulse,  see  Pulse  measurements 
resistance,  11-23 
reverberation,  12-48 


INDEX 


29 


Measurement,  transconductance,  4-12 
transmission,  11-32 
use  of  pulses,  9-02 
voltage,  11-17 
wire  line,  11-32 

Mechanical-acoustic  system,  5-66 
analogue,  5-59 
band-pass  filter,  11-63 
equivalent  of  heat,  1-79 
-fluid  system,  5-66 
impedance,  5-57,  6-03 
reactance,  5-59 
recordings,  13-37 

recording  disks,  13-41 

duplication,  13-41 

recording  instruments  used  in,  13-37 
recording  media,  13-40 
reproducing  instruments,  13-43 
magnetic  type,  13-43 
piezoelectric  crystal,  13-43 
reproducing  media,  13-40 
sources  of  distortion  in,  13-45 
resistance,  5-59 
scanning,  20-07 
systems,  5-56 

comparison  with  electrical  systems,  5-59 
energy,  5-57 
vibrations,  5-58 
units,  1-42 
Medical  applications  of  electricity,  23-01 

roeritgenography,  tubes,  4-86 
Medium-frequency  broadcasting,  use  of  direc- 
tional antennas  and,  6-74 
Megagram,  conversion  factors,  1-54 
Megaraeter,  conversion  factors,  1-47 
Megmho,  conversion  factors,  1-61 
Megohms,  conversion  factors,  1-61 
Meissner  oscillator,  7-84 
Mel,  12-17 

Melamine,  filled,  properties,  2-26 
-formaldehyde,  resins,  thermoaetting,  2-39 
glass  laminates,  2-39 
Melmac,  2-39 

Melting  point,  abbreviation,  1*71 
Mensuration,  1-17 
Mercury-arc  rectifiers,  4-77 
properties,  2-07 
-vapor  tube,  defined,  4-04 
Merit,  figure  of,  3-31 
Mesh  equations,  5-05 
Mesitylene,  dielectric  properties,  2-49 
Metal  gages,  1-66 
oxide-coated,  4-03 
tubes,  4-62 
Metalloids,  2-03 
Meter,  abbreviation,  1-71 
conversion  table,  1-47 
cubic,  abbreviation,  1-71 
conversion  table,  1-49 
dbm,  11-32 
-kilogram,  abbreviation,  1-71 

conversion  factors,  1-57 
-lambort,  14-17 
square,  abbreviation,  1-72 

conversion  table,  1-48 
standard  of,  1-42 
Methacrylate,  2-39 
properties,  2-28 

Methane,  dielectric  constant,  2-54 
Methyl  alcohol,  dielectric  properties,  2-49 
relation  between  dielectric  constant  and  resis- 
tivity, 2-51 

Metric  horsepower,  conversion  factors,  1-58 
multiples,  1-47,  1-48,  1-49,  1-54 


Metric  system,  1-42 

ton,  conversion  factors,  1-54 

waves,  1-80 

Wire  Gage,  1-69,  1-70 
Mho,  conversion  factors,  1-61,  1-62 
Mica,  2-39 

capacitors,  3-64 

cloth,  2-40 

glass-bonded,  2-38 

paper,  2-40 

pasted,  2-40 

plate,  2-40 

power  factor  at  high  frequencies,  2-34 

properties,  2-28 
Micabond,  2-40 
Micanite,  2-40 
Micarta,  2-40 

Microampere,  abbreviation,  1-71 
Microfarad,  abbreviation,  1-71 

conversion  factors,  1-62 
Micrograms,  conversion  factors,  1-54 
Microhenrys,  conversion  factors,  1-63 
Microhm-centimeters,  conversion  factors,  1-61 

-inches,  conversion  factors,  1-61 
Microhms,  conversion  factors,  1-61 
Micromho,  conversion  factors,  1-61,  1-62 
Micromicron,  abbreviation,  1-71 
Micron,  abbreviation,  1-71 

conversion  factors,  1-47 

square,  abbreviation,  1-72 
Microphone,  16-04 

carbon,  13-26 

choice  of,  16-05 

condenser,  13-24 

crystal,  13-25 

defined,  13-22 

directional  characteristics,  13-26 

force  on,  13-22 

graphical  symbol,  1-76 

location,  16-05 

magnetic-armature,  13-25 

moving-coil,  13-24 

moving-conductor,  13-23 

performance,  13-26 

placement,  16-05 

pressure,  13-22 

pressure-difference,  13-23 

ribbon,  13-23 

tests,  13-26 

types,  16-04 

Microvolts,  conversion  factors,  1-60 
Microwatt,  abbreviation,  1-71 
Microwave  antennas,  6-86 

instrument  landing  system,  22-28 

omnidirectional  radio  range,  22-30 
Microwaves,  1-80 

measurements,  11-69 

frequency,  11-84 
Middle  ear,  12-02 
Midohm,  properties,  2-07 
Miles,  conversion  table,  1-47 

nautical,  conversion  factors,  1-47 

per  hour,  abbreviation,  1-72 

square,  conversion  table,  1-48 

statute,  conversion  factors,  1-47 
Military  pace,  conversion  factors,  1-47 
Miller  capacitance  effect  of  wide-band  amplifier 
tubes,  7-43 

effect,  4-18,  5-49 
Milliampere,  abbreviation,  1-71 
Milligram,  abbreviation,  1-71 

conversion  factors,  1-54 
Millihenry,  abbreviation,  1-71 

conversion  factors,  1-63 


30 


INDEX 


Milliliter,  abbreviation,  1-72 

conversion  factors,  1-49 
Millimeter,  abbreviation,  1-72 
conversion  table,  1-47 
gage,  2-13 

square,  conversion  table,  1-48 
Millimetric  waves,  1-30 
Millimicron,  abbreviation,  1-72 
Millivolt,  abbreviation,  1-72 

conversion  factorst  1-60 
Mils,  circular,  abbreviation,  1-71 

conversion  factors,  1-48 
conversion  table,  1-47 
square,  conversion  factors,  1-48 
Mineral  oil,  2-52 

dielectric  properties,  2-49 
Minerallac,  2-40 

Miner's  inch,  conversion  table,  1-52 
Minima,  1-13 

Minims,  conversion  factors,  1-50 
Minimum,  abbreviation,  1-72 
attenuation,  5-12,  6-57 
audible  field  in  hearing,  12-05 
audible  pressure  in  hearing,  12-05 
conductance,  5-08 
loss,  5-12 
phase,  5-12 
reactance,  5-08 
resistance,  5-08 
resolvable  solid  angle,  20-02 
susceptance,  5-08 
visible,  14-43 
Minor  third,  10-09 
Minute,  abbreviation,  1-72 
Minute  (angular  measure),  abbreviation,  1-72 
Minutes  (angle),  conversion  table,  1-51 
Minutes  (time),  conversion  table,  1-51 
Mixed  highs,  20-66 
Mixer  matching  network,  16-08 

volume  controls,  16-07 
Mixing  of  sound,  16-05 
Mks  system,  1-42,  1-45 

rationalized,  1-44,  1-45,  1-46 
unrationalized,  1-44 
Mode  separation,  magnetrons,  4-45 
Modes,  4-54,  7-101 

cavity  resonators,  7-101 
quality  factor,  7-101 
Moding,  magnetrons,  4-45 
Modulated  amplifiers,  7-75 
Modulation,  amplitude,  7-71 
carrier  wave,  7-70 
cavity,  in  speech,  12-19 
characteristics  of  pulses  and,  9-23 
defined,  7-70 
distortion,  4-24 
frequency,  see  F-m 
frictional,  in  speech,  12-19 
grid,  7-73 

Heising  amplitude,  7-74 
index  of  frequency  modulation,  8-03 
modulating  wave  of,  7-70 
percentage,  11-61 
phase,  see  Phase  modulation 
plate,  7-74 
polarity,  20-19 
power  and  efficiency   of   grid-bias-modulated 

amplifiers,  7-73 
products,  11-62 
pulse,  see  Pulse  modulation 
start-stop,  in  speech,  12-19,  12-20 
systems,  7-72 

comparison,  7-75 
transformer,  6-19 


Modulation,  types,  7-71 

amplitude  modulation,  7-71 
frequency  modulation,  7-71 
vocal-cord,  in  speech,  12-19 
Modulator-amplifier  coupling  circuits,  7-74 
Modulators,  7-70 
pulse-time,  9-23 
thyratron,  9-22 
vacuum-tube,  9-22 
Modulus  (of  complex  number),  1-06 

of  elasticity,  2-26 
Molded  compounds,  2-40 
Molybdenum-iron  alloys,  2-62 
permalloy,  2-62 
properties,  2-07,  2-10 
Moment,  electric,  symbol,  1-73 
magnetic,  symbol,  1-73 
of  force,  conx-ersion  table,  1-56 
of  inertia,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 
Monaural  hearing,  16-02 

minimum  audible  pressure  (hearing),  12-06 
Mond,  properties,  2-07 
Monel  metal,  properties,  2-07 
Monitoring  facilities,  10-09,  16-26 
Monochord,  11-12 
Monochrome  television,  20-02 
Monocular  television,  20-02 
Monoscope,  15-25 

Months  (average),  conversion  table,  1-51 
Morse  code,  18-03 
Motion,  perception  of,  14-44 

-picture  studios,  acoustic  design,  12-41 
Motional  impedance  circle,  T>-66 

joints  of  wavo  guidon,  10-20 
Motor  noise,  11-64 
Motorboating  of  amplifiers,  7-04 
Moving-coil  microphono,  13-24 
-coil  speaker,  13-11 
-conductor  microphone,  13-23 
-conductor  speaker,  13-11 
-conductor  telephone  receiver,  13-17 
object,  resolution  of,  20-03 
MR  Resins,  2-34 
Mu  factor,  4-13 
defined,  4-06 

measurement  cireuit,  4-13 
Multianode  tube,  defined,  4-75 
Multiband  sets  (radio),  i-f  amplifiers  and,  7-58 
Multi-electrode  tubes,  5-45 
Multiform  glass,  2-40 
Multigrid  tubes,  4-03 
Multipath  reception,  distortion  due  to,  8-29 

of  f-m  signal,  8-29 
transmission,  causing  frequency  modulation, 

8-26 

downward  amplitude  modulation  and,  8-29 
selective  fading  and,  8-29 
Multiple  echo,  12-40 
product,  1-02 
-pulse  generator,  9-24 
track  radar  range,  22-30 
tuned  antenna,  6-80 
tuning  positions  of  f-m  receivers,  8-17 
-unit  tube,  defined,  4-04 
Multiples,  defined,  22-05 
Multipliers,  frequency,  9-13 
Multistage  amplifier,  7-03 
Multivibrators,  free-running,  9-18 
relaxation  oscillation  in,  7-84 
repetition  rate  of  pulses  determined  by,  9-19 
triggered,  9-18 
Mumetal,  2-62 


INDEX 


31 


Musa  receiving  antennas,  6-83 
Muscovite,  2-39,  2-40 
Muscular  reaction,  23-03 
Music,  12-24 

audible  frequency  range  of,  12-30 

effect  of  cutoff  frequency  on  orchestral  duality. 
12-30 

effects  of  distortion,  12-29 

optimal  reverberation  times,  12-75 
at  different  frequencies,  12-75 

peak  power,  12-24 

rooms,  acoustic  design,  12-41,  12-74 
noise  levels  acceptable  in,  12-58 
sound  insulation,  12-57 

scales,  11-09 

transmission  of,  teats  for,  12-27,  12-28 
Musical  instruments,  powers  produced  by,  12-24 
Mutual  capacitance,  11-24 

characteristic,  5-41 

conductance,  4-12 
of  tubes,  6-11 

impedances,  6-07 
in  coupled  circuits,  6-07 

inductance,  1-73,  6-07,  6-13 
Mycalex,  2-38,  2-40 

K,  2-38 

Mycroy,  2-38,  2-40 
Myriagram,  conversion  factors,  1-54 
Myriameter,  conversion  factors,  1-47 

square,  conversion  factors,  1-48 
Myriametric  waves,  1-80 

n  factorial,  1-02 
Napierian  logarithms,  1-19 
Natural  frequency  of  the  ear,  12-04 

logarithm,  abbreviation,  1-71 

logarithms,  1-19 
Nautical  measure,  conversion  factors,  1-47 

miles,  conversion  factors,  1-47 
Navagllde  instrument  landing  system,  22-28 
Navaglobe,  22-31 

Naval  Observatory  time  signals,  18-40 
Navar,  22-29 
Navascope,  22-29 

Navigation,  see  also  Air  navigation;  Marine  navi- 
gation 

aide,  22-04 

Navy  announce  equipment,  16-18 
NBS  gage,  1-69 
Negative  condenser,  5-59 

feedback  amplifiers,  7-31,  7-51 
effect  on  distortion,  7-52 

glow,  4-50 

modulation,  20-19 

roaiwtance  magnetrons,  4-40 
Neoprene,  2-40 

properties,  2-28 
Neper,  1-37 

Nernst  filament,  properties,  2-10 
Nerve  conduction,  12-03 

fibor,  12-04 

Net  ton,  conversion  factors,  1-54 
Network  switching  equipment,  for  program  dis- 
tribution, 16-27 
Networks,  all-pass,  5-21 

behavior,  5-29 

bridge,  6-13 
inverse,  5-09 

complementary  impedances,  5-10 

connective,  IOMH  equalizers,  5-16,  5-18 
losN-phuHO  relation,  5-17 
phase  equnliflerB,  5-16 

corrective,  5-16 

cutoff  frequencies,  6-38 


Networks,  delay,  5-30 
differentiating,  5-29 
driving  point  impedances,  6-35 
excess-phase,  5-33,  5-35 
four-terminal,  see  Four-terminal  networks 
general  filter,  6-33 
having  any  prescribed  passive  transfer  function, 

5-12 

image  impedance,  6-37 
image  transfer  functions,  6-37 
impedance  function,  6-35 
integrating,  5-29 

inverse  or  reciprocal  impedances,  5-09 
lattice,  5-13 

linear  passive,  see  Linear  passive  networks 
L-type,  5-13 
mesh  for,  6-34 
method  of  obtaining  second  image  impedance, 

6-40 

minimum-phase,  5-33 
multi-mesh,  5-05 
nodal  equations,  6-34 
of  pure  reactances,  5-08 
of  resistances  and  capacitances,  5-09 
of  resistances  and  inductances,  5-09 
oscillatory,  transient  response,  5-31 
parallel  type,  6-35,  6-36 
phase-correcting,  5-33 
reactance  function,  6-35 
reactive,  5-08 
ladder-type,  5-08 
Mayer's  theorem,  6-61 
recurrent,  5-22 

uniform  lines,  5-24 
repeating.  5-30 
resonant,  5-30 
series-type,  6-35,  6-36 
short-  and  open-circuit,  6-37 
simple,  transient  response,  5-31 
symmetrical,  5-23 

current  and  voltage  relations,  5-23 
impedance  relations,  5-23 
incident  and  reflected  waves,  5-23 
tandem  combination,  5-23 
T  and  ir,  5-13 
theorems,  5-12 

compensation,  5-12 
reciprocity,  512 
TheVenin's,  5-12 
transients  in,  5-26 

superposition  theorem,  5-34 
two-terminal,  inverse,  5-09 

reactive,  6-35 

with  distributed  constants,  5-24 
Neural  pulses,  12-11 

conduction  of,  to  brain,  12-04 
Neuritis,  23-06 
Neutralization,  7-29 
Neutralized  receiver,  7-119 
Neutralizing  transformer,  10-94 
New  British  Standard  Wire  Gage,  1-69 
Newton-meter,  conversion  factors,  1-56 
Nichrome,  properties,  2-07 
V,  properties,  2-07,  2-10 
Nickel,  2-03 

-chromium,  properties,  2-07 
-iron  alloys,  2-62 
properties,  2-07 
-silver,  properties,  2-07 
steel,  properties,  2-07 
Nickelin,  properties,  2-07 
Nicraloy,  properties,  2-07 
Nilvar,  properties,  2-07 
Nirex,  properties,  2-07 


32 


INDEX 


Nitrobenzene,  dielectric  properties,  2-50 
Nitrogen,  dielectric  constant,  2-54 
dielectric  properties  of  liquid,  2-50 
minimum  sparking  potentials,  2-54 
Nitron,  2-36,  2-41  j 
Nixonite,  2-41 
Nixonoid,  2-36,  2-41 
Nodal  analysis,  method,  5-06 
equations,  5-06 

for  networks,  6-34 
Nodes,  current,  5-26 

voltage,  5-26 

Noise  (noises),  abatement,  12-57 
allowable  residual,  16-02 
analysis,  examples  of,  12-58 
analysis  of  small  synchronous  motor,  11-64 
atmospheric,  10-42 
audible  frequency  range,  12-30 
currents,  4-20 

effects  on  articulation,  12-33 
extraneous,  effects  on  articulation,  12-33 
frequency  induction  of,  10-74 
generated  in  vacuum  tubes,  4-20 
in  buildings,  sound  levels  of,  12-59 
inductive  coupling,  10-78 
inductive  influence,  10-75 
inductive  mitigation  of,  10-82 

coupling  factors,  10-82,  10-83 

influencing  factors,  10-82 

susceptiveness  factors,  10-82,  10-85 
inductive  ausceptiveness,  10-80  ' 

levels  acceptable  in  different  buildings,  12-58 
measurement,  11-36,  11-44,  12-57 
out-of-doors,  sound  levels  of,  12-59 
positive-ion,  4-22 
random,  7-127 
range,  12-30 

ratio  of  speech  levels  to,  12-72 
-Deduction  factors,  12-69 

ceilings  and,  12-69 

for  different  amounts  of  noise,  12-69 

walls  and,  12-69 

windows  and,  12-69 
-reduction  system,  13-49 
resistor,  3-13 

sound  insulation  against,  12-57 
sound-level  meters  in  measurement,  12-58 
sound  levels  in  or  near  buildings,  12-59 
spectrum,  11-59 
suppression,  7-126 
thermal-acoustic,  12-06 
traffic,  analysis,  example  of,  12-58 

sound  insulation  against,  12-57,  12-58 
tube,  4-23 
Non-linear  circuits,  5-37,  7-121 

approximate  series  expansion  for  plate  cur- 
rent of  a  triode,  5-41 

capacitance  variation,  5-41 

characteristics  of  triode  with  load,  5-42 

current-voltage  characteristic,  5-38 

harmonic  analysis  of  current  for  a  sinusoidal 
applied  voltage,  5-4,6 

inductance  variation,  5-40 

multi-electrode  tubes,  analyses  for,  5-45 

power  series  solution,  5-38 

solution,  5-38 

successive  approximations,  5-45 

trigonometric  series,  5-39 
distortion,  5-38 

effect  of,  12-34 
phase  characteristics  causing  f-m  distortion, 

8-26 

Non-loaded  cable  circuit,  10-03 
open-wire  lines,  10-03 


Non-magnetic  materials,  2-64 

Non-matched  impedances,  in  single-mesh  circuits, 

6-06 

Non-oscillating  magnetrons,  4-40 
Non-polar  materials,  2-22 
Non-uniform  delay,  f-m  distortion  and,  8-20 
Normal  black  level,  20-20 

induction,  defined,  2-60 

modes  of  electromagnetic  fields,  7-95 

permeability,  defined ,  2-60 
Nose  cavities,  speech  and,  12-19 
Null  in  transmission,  f-m  distortion  and,  8-25) 
Number  of  conductors  or  turns,  symbol,  1-73 

of  phases,  symbol,  1-73 

of  poles,  symbol,  1-73 
Numeric,  1-46 

Numerical  pitch  scale,  12-17 
Nylon,  2-41 

properties,  2-28 

Oboe,  12-30 

Observation,  errors,  1-15 

Observations,  probable  value  of  several,  1-15 

weighted,  1-15 

Octane,  dielectric  properties,  2-50 
Octave,  11-09 
Octode,  defined,  4-04 
Odd-line  interlace,  20-06 
Oersted,  1-46 

conversion  factors,  1-64 
Office  buildings,  wound  insulation,  12-57 
Offices,  private,  noi.se  hwdn  acceptable  in,  12-58 

public,  noise  lovol  aeooptablo  in,  12-58 
Ohm,  1-43,  1-4-1,  1-45,  1-46 

abbreviation,  1  -72 

-centimeter,  defined,  2-02 

conversion  factors,  1-0 1 

-inch,  defined,  2-02 

-motors,  conversion  factors,  1-61 

per  centimeter  cube,  defined,  2-02 

per  inetor-gram,  1-61 

per  mil  foot,  1-61 
defined,  2-02 

per  unit  weight,  2-16 

thermal,  1-65 
Ohmax,  properties,  2-07 
Oils,  insulating,  2-52 
Old  English  Wire  Gage,  1-69, 1-70 
Olive  oil,  dielectric  properties,  2-50 
Omnidirectional  antennas,  using  vertical  polari- 
zation, 6-84 

range,  22-18 

One-shot  amplifiers,  7-31,  7-53 
Open-wire  circuit,  10-03 
Opens,  11-41 
Operating  conditions,  changing,  4-26 

range,  4-06 

Operational  impedance,  5-43 
Operator,  differential,  5-29 

integral,  5-29 
Optical  system,  depth  of  field,  14-13 

depth  of  foous,  14-13 

electron,  general  theorems,  14-63 

light-valvo  recording,  13-48 

reflective,  for  television  projection,  14-20 

stops,  14-13 
Optics,  14-02 

electron,  14-49 

geometrical,  14-02 
Format's  principle,  14-02 

vision,  14-25 

Optimal  reverberation  times,  12-75,  16-03,  16-11 
Optimum  angle  of  current  flow,  7-131 

horn,  6-79 


INDEX 


33 


Orchestras,  12-24 

quality,  12-31 
Order  of  modulation,  11-56 
Organs  of  speech,  12-19 
Orthicon,  15-26 

images,  see  Imago  orthicon 

resolution,  15-27 

sensitivity,  15-27  ' 

siRnal-to-noise  ratio,  15-27 

uniformity,  15-27 
Oscillating  joints  of  wave  guides,  10-26 

magnetrons,  4-40 
Oscillations,  5-30 

blocking  7-86 

condition  for  persintence,  7-84 

conditions  for  self-oscillation,  7-84 

non-linear  theory,  7-83 

of  Ras-filled  tubes,  7-91 

oscillatory  circuit  design,  7-86,  7-87 

paranitic,  in  amplifiers,  7-28,  7-29 

prevention  of,  in  amplifiers,  7-29 

relaxation,  7-84,  7-86 

theory  of,  7-83 

van  der  Pol's  equation  for  non-linear  theory, 
7-85 

variation  in  frequency  of,  in  oscillator  circuits, 

7-94 

Oscillator,    7-83,    see   also    Oscillatory   circuits; 
Vacuum-tube  oscillators 

at  hitfh  frequencies,  7-89 

audio-froquonny,  11-89 

BarkhauHori,  7-91 

buffer  amplifier,  4-52 

circuilH,  conventional,  7-84 

dUNHifirtations,  7-83 

Ooipittn,   conytant   frequency  circuit  derived 
from,  7-88 

corxHtant-frotiuonoy,  7-87 

wyHtnl,  7-92 

do/mad,  7-83 

eliBtortionloHH,  7-90 

etactrutnoohanittal,  7-91 

equivalent  circuit,  9-23 

Kan-filled,  deiontaation  time  of  gas  in  operation, 

7-91 
ionization  time  of  gas  in  operation,  7-91 

Hartley,  7-93 

hetesrodyno,  1 1-31 

interpolating,  11-10 

frequency  compariwon  with,  11-10 

magneton  trietion,  7-92 

piezoelectric,  crystal,  7-92 

plate  modulation  of,  and  cla««  C  amplifiers,  7-85 

pulne-modulated  carrier-frequency,  9-22 

pul«o  modulation,  9-21 

quarts  crystal,  7-92 

r-f,  11-00 

roHifitanca-eapacitance  tuned,  7-90 

Holf-cxmtnd,  4-49 

separately  excited,  7-88 

niuuiioUial,  7-K6 

Bpark-ftup,  7-94 

synchronization,  7-88 

tracking  of  Hiiperhoterodyne  receiver,  7-121 

triodo,  7-89 

tuned-plato  tuned-grid,  7-83 

timing-fork,  7-91 

-typo    reiayw,    elootromatfnetioally    operated, 

21-23 
ol«d.roHtati(!ttIly  operated,  21-23 

UHo  of  quartz,  7-92 

two  of  Rochollo  «alfc,  7-92 

une  of  totirmalino,  7-92 

velocity  variation,  11-92 


Oscillatory  circuits,  design,  7-86,  7-87 

ordinary,  resonant  concentric  lines  in,  7-SP 

transients,  5-27 
Oscillograph  tubes,  multiple  gun,  15-47 

-type  cathode-ray  tubes,  see  Cathode-ray  tuoes, 

oscillograph-type 
Oscilloscopes,  cathode-ray,  2-76 

pulse  measurements  by,  9-10 

synchroscope  type,  9-12 
Osmium,  properties,  2-07 
Ossicles,  12-02 

Ostwald  calorie,  conversion  factors,  1-57 
Ounce,  abbreviation,  1-72 

conversion  factors,  1-54,  1-55 

fluid,  conversion  factors,  1-49,  1-50 

-foot,  abbreviation,  1-72 

-inch,  abbreviation,  1-72 
Outdoor  address  systems,  16-15 
Outer  ear,  12-02 
Output  capacitance,  denned,  4-06 

gap,  4-51 

power,  standardization,  11-96 

transformers,  6-17 

wave  spectrum,  1 1-56 

Overall  selectivity  curve,  in  coupled  circuits,  6-10 
Overbunching,  4-53 
Owen  bridge,  11-28 
Oxide-coated  metal,  4-03 
Oxygen,  dielectric  constant,  2-54 

minimum  sparking  potentials,  2-54 
Ozite,  15-54 
Ozokerite,  2-41 

properties,  2-28 

Pace,  military,  conversion  factors,  1-47 
Pad,  6-05 

-type  resistors,  3-09 
Padding  condenser,  3-56 
Paging  systems,  16-16 
Painting  of  acoustic  material,  12-57 
Paired  echoes,  5-33 
Palladium,  properties,  2-07 
Palm,  conversion  factors,  1-47 
Pancake  winding,  6-24 
Panel  dial  telephone  system,  17-17 

decoder,  17-20 

dial  pulse  register  circuit,  17-20 

operation,  17-23 

panel-type  selector,  17-17 

sequence  switch,  17-20 
Panelyte,  2-41 
Paper,  insulating,  2-41 

kraft,  properties,  2-28 
Parabola,  cutoff,  4-40 

equation,  1-05 

mensuration,  1-17 

Parabolic  reflectors  of  antennas,  6-78 
Paraboloid  cylinders  of  reflectors  of  antennas, 
6-78 

mensuration,  1-18 
Paraboloidal  antenna,  6-63 
Paraffin,  properties,  2-28 

waxr  relation  between  dielectric  constant  and 

resistivity,  2-51 
Paragonite,  2-39 
Paragutta,  2-41 

Parallel  resonant  circuits,  6-04 
tuning,  6-05 

type  networks,  6-35,  6-36 
Parallelogram,  mensuration,  1-17 
Paramagnetic  materials,  2-57 
Parameters,  image,  see  Image  parameters 
Parasitic  oscillation  in  amplifiers,  7-28,  7-29 
Paraxial  rays,  denned,  14-07 


34 


INDEX 


Partitions,  rigid,  coefficients  of  sound  transmis- 
sion, 12-65,  12-66 
Parts,  integration  by,  1-12 
Pass  band  of  filters,  6-33 

image  impedance  and,  6-36 

image  transfer  constant,  6-36 
Passive  circuits,  attenuators,  6-05 

elements,  6-02 

pads,  6-05 

Pattern  sharpaess,  defined,  22-05 
PBX  boards,  see  Private  branch  exchange  (PBX) 

boards 
Peak  angle  swing,  frequency  modulation  and,  8-06 

detector,  7-79 

(or  crest)  forward  anode  voltage,  defined,  4-05 

(or  crest)  inverse  anode  voltage,  denned,  4-05 

power  measurements,  11-81 

speech  power,  12-22 

voltmeter,  11-17 

Peanut  oil,  dielectric  properties,  2-50 
Pecks,  conversion  factors,  1-50 
Peerless,  properties,  2-07 
Pennyweights,  conversion  factors,  1-54 
Pentanes,  conversion  factors,  1-65 

dielectric  properties,  2-50 
Pentode,  characteristics,  4-37,  4-38,  4-39 

defined,  4-04 

graphical  symbol,  1-77 ; 

power  amplifiers,  7-13 

suppressor-grid,  4-03 

type  tubes,  6-11 

voltage  amplifiers,  7-11 
Perbunan,  2-35,  2-41 
Percentage  articulation  for  rooms,  12-69 

modulation,  measurement,  11-61 
Perception,  of  depth,  14-45 

of  motion,  14-44 

temporal  aspects,  14-33 
Perch,  conversion  factors,  1-47 

of  stone,  conversion  factors,  1-49 
Period,  hyperbolic,  1-10 

symbol,  1-73 

transient,  5-27 

trigonometric,  1-08 

Periodic  waves,  frequency  spectrum,  5-28 
Permalloy,  2-62,  2-70 

powdered,  2-62 
Permanent-magnet  materials,  typical  properties, 

2-67 
Permeability,  2-57 

a-c,  defined,  2-60 

curves,  2-57 

incremental,  defined,  2-60 

initial,  defined,  2-60 

magnetic,  symbol,  1-73 

normal,  defined,  2-60 

relative,  cgs  unit,  1-46 
mks  unit,-46 
symbol,  1-46,  1-73 

reversible,  2-59 
defined,  2-60 

space,  1-44 
cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 

superposed,  2-59 

tuning,  3-52 
Permeameters,  2-74 
Permeance,  symbol,  1-73 
Permendur,  2-62 
Permenorm,  2-62 
Perminvars,  2-62 
Permittivity,  1-73 
Permutations,  1-03 


Petrolatum,  2-41 

properties,  2-28 
Petroleum,  dielectric  properties,  2-50 

ether,  dielectric  properties,  2-50 

oils,  relation  between  dielectric  constant  and 

resistivity,  2-51 
Phantom  circuit,  10-03 

transposition,  10-78 
Pharynx,  speech  sounds  and,  12-20 
Phase  angle,  of  resistor,  1 1-24 
symbol,  1-73 

constant,  symbol,  1-73 

delay,  5-36 

difference,  2-22 

distortion,  20-05 

linear,  5-36 

modulation,  see  also  Frequency  modulation 
discriminator  action  of  detectors,  8-03,  8-19 
Phasitron  tube  used  in,  8-11 
quasi-mechanical  method,  8-11 
sideband  distribution  for,  8-06 
systems,  8-02 
voltage  vector  with,  8-03 

modulators  used  for  emergency  transmitters, 
8-15 

propagation,  velocity  of,  5-25 

shift,  insertion,  5-16 
minimum,  5-17 

slope,  5-36 

swing  measurement  of  f-m  equipment,  8-08 

velocity,  5-36 

Phasing  of  speakers,  16-14 
Phasitron  tube,  frequency  modulation  using,  8-1 1 
Phenix,  properties,  2-07 
Phenol,  dielectric  properties,  2-50 

fiber,  2-41 
Phenolic  cast,  properties,  2-28 

glass  base,  power  factor  at  high  frequencies, 
2-34 

insulating  materials,  2-41 

laminates,  properties,  2-28 

mica  filled,  power  factor  at  high  frequencies, 
2-34 

moldings,  properties,  2-28 

paper  base,  power  factor  at  high  frequencies, 
2-34 

resins,  2-41 
Phenolite,  2-41 
Phi-phenomenon,  14-44 
Phlogopite,  2-39,  2-41 
Phon  of  loudness  levels  (sound),  12-11 
Phonetic  printing,  visible  speech  and,  12*21 

speech  power,  12-22 
Phonograph,  distortion,  12-35 
Phosphor-bronze,  properties,  2-07 

crystals,  luminescence-active  centers  in,  types, 

15-36 

Phosphorescence,  defined,  15-30 
Phosphors,  characteristics,  15-37 

corpuscular  excitation,  15-34 

defined,  15-31 

mechanisms,  15-34 

photon  excitation,  15-35 

preparation  and  notation,  15-32 
Phot,  1-46 

Photocells,  barrier,  15-13 
Photoconductive  cells,  15-11 

amplification,  15-13 

frequency  response,  15-12 

sensitivity,  15-12 

wavelength  response,  15-11 
Photoelectric  cathode,  graphical  symbol,  1-77 

cell,  16-22 

relays,  21-21 


INDEX 


35 


Photoelectric  tube,  15-02 
Photoemissive  cells,  15-04 
amplification,  15-10 
gas-filled,  15-06 

measuring  circuits  for  use  with,  15-09 
sensitivity,  15-07 
vacuum,  15-06 
Photofluorograph,  23-16 
Photographic  sound  recordings,  13-47 
flash-lamp  system,  13-51 
Kerr  cell  system,  13-52 
light- valve  system,  13-48 
noise-reduction  system,  13-49 
reflecting-galvanometer  system,  13-50 
requirements,  13-54 
sound-on-film  system,  13-52 
sound  tracks,  variable-area,  13-53 

variable-denisty,  13-53 
transparencies,  20-03 
Photoluminescence,  defined,  15-30 
Photometry,  14-14 
illuminance,  14-16 
luminance,  14-16 
luminous  efficiency  in,  14-15 
luminous  emittanoe,  14-17 
luminous  intensity  in,  14-15 
physical,  14-17 

relations  in  non-visual  optical  systems,  14-18 
units  in,  14-17 
visual,  14-17 

Photomultiplier  tube,  15-10 
Photon  excitation,  of  phosphors,  15-35 
Photoresponsive  devices,  15-02 
classification,  15-02 
photoelectric,  15-02,  15-16 
thermal,  15-02,  15-03 
Phototubes,  4-02 
defined,  4-04 

graphical  symbol,  1-76,  1-77 
Photovoltaic  cells,  characteristics,  15-15 
Physical  constants  and  ratios,  1-79 
Physiological  emf,  23-08 

requirements,  television,  20-02 
Piano,  12-25,  12-30 
Piccolastic,  2-41 
Piccolo,  12-25,  12-30 
Pick-up  tubes,  application,  15-29,  20-07 

requirements,  15-19,  20-30 
Picture  display  devices,  20-08 
receivers,  block  diagram,  9-07 
television,  20-03 
transmission,  by  pulses,  9-06 
transmitters,  block  diagram,  9-07 
scanning  functions,  9-07 
synchronizing  functions,  9-07 
tubes,  defined,  15-46 

deflection  and  focus,  15-47 
Piezoelectric  crystal  oscillator  circuits,  7-93 
crystal  oscillators,  7-92 
crystals,  13-55 
application,  13-68 
defined,  13-55 
definition  of  effects,  13-56 
microphone,  13-25 
plate,  graphical  symbol,  1-76 
properties  of  crystal  in  oscillators,  7-93 
telephone  receiver,  13-18 
Pigmentation,  23-07 
Pinna  of  ear,  12-02 
Pint,  abbreviation,  1-72 

conversion  factors,  1-49,  1-50 
Pipe  organ,  12-25 
Pistons,  13-04 

enclosed  back,  13-05 


Pistons,  multiple,  13-04 

Pitch,  2-35 

Pitch  (sounds),  change,  12-16 

comparisons,  12-16 

half,  judgments  of,  12-16 

intervals,  12-17 

numerical  scale,  12-17 

of  low-frequency  tones,  auditory  nerves  and, 
12-03 

variation,  12-16 
Placement  of  microphone,  16-05 

of  speaker,  16-13 
Placet,  properties,  2-07 
Planck  constant,  1-79 
Planck's  radiation  law,  15-30 
Plane  angle,  conversion  table,  1-51 

of  incidence,  5-53 

waves,  progressive,  5-51 
Plant  broadcast  systems,  16-16 
Plaskon,  2-41 

Melamine,  2-39 
Plastacele,  2-36,  2-41 
Plaster,  12-54,  12-55,  12-66 

board,  12-65 
Plate  characteristic,  4-11 

current  detector,  resistance  coupled  to  succeed- 
ing amplifier  tube,  7-78 

currents,  denned,  4-05 
of  a  triode,  5-44 

denned,  4-04 

efficiency,  7-09 

forms  of  capacitors,  3-57 

graphical  symbol,  1-77 

-grid  transconductance,  5-43 

modulation,  7-74 

piezoelectric,  graphical  symbol,  1-76 

power,  of  receivers,  7-106 

resistance,  4-11 

transformer,  6-26 

voltage,  defined,  4-05 
Platinite,  properties,  2-07 
Platinoid,  properties,  2-07 
Platinum-cobalt  alloys,  2-68 

-iridium,  properties,  2-08 

properties,  2-07,  2-08 

-rhodium,  properties,  2-08 
Plai,  2-41,  2-42 
Pleurisy,  23-06 
Pleriglas,  2-39,  2-41,  2-47 
Pliolite,  2-41,  2-43 

Plosive  release  in  visible  speech,  12-21 
Plug-in  type  resistors,  3-09 
Pneumatic  speaker,  13-17 
Point  source  of  radiation,  13-03 
Polar  diagram,  4-49 

materials,  2-22 

Polarity  of  modulation,  20-19 
Polarization,  electric,  symbol,  1-73 

magnetic,  symbol,  1-73 

television,  20-20 
Pole,  conversion  factors,  1-47 

lines,  10-49 

basic  conductor  loadings,  10^51 
guying,  10-55 
joint  use,  10-50 
loading  calculations,  wire  equivalents,  10-52 

strength,  magnetic,  cgs  unit,  1-46 
TnVs  unit,  1-46 
symbol,  1-46,  1-73 
Polectron,  2-41,  2-47 
Poles,  cross-arms  on,  10-55 

open  wire,  10-55 

spacing,  10-55 

treatment,  10-54 


36 


INDEX 


Police  radio,  16-35 
equipment,  16-37 
frequency  allocation,  16-36 
power  and  range  of,  16-37 
Polyamides,  2-41 

Polychlorostyrene,  power  factor  at  high  frequen- 
cies, 2-34 

Polycylindrical  sound  diffusers,  12-70,  12-71 
Polydichlorostyrene,  2-42 

properties,  2-28 
Polyethylene,  2-42 

power  factor  at  high  frequencies,  2-34 
properties,  2-28 
Polyflex,  2-42 
Polystyrene,  2-34,  2-42 
modified,  2-42 

power  factor  at  high  frequencies,  2-34 
properties,  2-28 
Polytetrafluoroethylene,  2-42 

power  factor  at  high  frequencies,  2-34 
properties,  2-28 
Polythene,  2-42,  2-47 
Polyvinyl  acetal  resin,  2-34 

carbazole,  power  factor  at  high  frequencies, 

2-34 

Poncelet,  conversion  factors,  1-58 
Pool  cathode,  denned,  4-75 
graphical  symbol,  1-78 
-cathode  tubes,  4-75 
available,  4-76 
classification,  4-77 
tube,  denned,  4-75 
Porcelain,  2-42 

properties,  2-10,  2-28 

wet  process,  power  factor  at  high  frequencies, 

2-34 

Porches,  television,  20-16 
Porous  materials,  12-62 
Positive-ion  current,  4-03 
modulation,  20-19 
noise,  4-22 

Potassium,  properties,  2-08 
Potential  difference,  1-60 
electric,  symbol,  1-73 
gradient,  cgs  unit,  1-46 
conversion  table,  1-60 
inks  unit,  1-46 
symbol,  1-46 
'  magnetic,  symbol,  1-73 
magnetic  vector  p.,  symbol,  1-73 
retarded  scalar,  symbol,  1-73 
retarded  vector  p.,  symbol,  1-73 
Potentiometers,  3-17 

carbon  composition  type,  3-20 
defined,  3-02 
step-type,  3-20 
wire-wound,  3-18 
Pound,  abbreviation,  1-72 
British  imperial,  1-42 
conversion  factors,  1-54,  1-55 
-foot,  abbreviation,  1-72 

conversion  factors,  1-56 
-inch,  abbreviation,  1-72 
per  square  foot,  abbreviation,  1-72 
per  square  inch,  abbreviation,  1-72 
XT.  S.  avoirdupois,  1-42 
Poundals,  conversion  factors,  1-55 
Power,  active,  cgs  unit,  1-46 
inks  unit,  1-46 
symbol,  1-46,  1-73 
amplification,  7-03 
amplifiers,  7-131 

Doherty,  7-131,  7-132 

negative  feedback  applied  to,  7-133 


Power  amplifiers,  plate-circuit  modulation  used 

in,  7-132 

r-f  harmonic  radiation,  7-132 
shunt  neutralization  employed  in,  7-132 
apparent,  symbol,  1-73 
circuit  transposition,  10-79 
conversion  table,  1-58 
factor,  2-22 
abbreviation,  1-72 
symbol,  1-73 

gain  of  receiving  antennas,  6-72 
level,  1-41 
loss,  2-22 

measurement,  11-102 
musical  instruments,  12-24 
pulse,  9-10 
radiated,  5-52 
reactive,  cgs  unit,  1-46 

mks  unit,  1-46 
'     symbol,  1-46,  1-73 
reference  levels,  1-41 
requirements,  16-13 
series  solution,  5-38 
speech,  12-22 
supply,  7-106 
systems,  coordination  of  communication  and, 

10-67 

transfer,  5-15 
insertion  loss,  5-15 
insertion  phase  shift,  5-16 
reflection  factor,  5-15 
symmetry  factor,  5-15 
transition  loss,  5-15 
transformers,  6-26 

calculation  of  performance,  6-29 
construction,  6-27 
copper  loss  of  windings,  6-29 
core  lows,  6-28,  6-29 
design  procedure,  6-28 
efficiency,  defined,  6-30 
flux  density,  6-28 
heating,  6-29 
insulation,  6-29 

regulation  of  secondary  winding,  6-30 
size,  6-26 

volt-ampere  rating,  6-26 
-type  resistors,  layer  windings,  3-07 

non-inductive  windings,  3-07 
Poynting's  vector,  5-51 

symbol,  1-73 

Practical  electrical  units,  1-44 
Precision-type  resistors,  3-07 
Preferred  resistance  values,  3-12 
Premier,  properties,  2-08 
Preselector,  7-119 
Pressboard,  2-43 
properties,  2-28 
Pressure,  audible,  12-05 
cgs  unit,  1-46 
conversion  table,  1-56 
difference  microphone,  13-23 
mks  unit,  1-46 
symbol,  1-46 
wind,  10-53 
Prestite,  2-43 

properties,  2-28 

Primary  daytime  coverage,  16-32 
electrical  quantities,  11-16 
feed  of  microwave  antennas,  6-77 
Principal  diagonal  (determinant),  1-04 
night-time  coverage,  16-32 
service,  16-32 
Prisms,  electron,  14-62 


INDEX 


37 


Prisms,  with  parallel  sides  and  parallel  ends,  men- 
suration, 1-18 
Private   branch   exchange   (PBX)  boards,   dial 

units,  17-116 
equipment,  17-114 
manually  operated,  17-114 
protection,  17-116 
Probable  error,  in  result  calculated  from  means  of 

several  observed  quantities,  1-16 
of  any  one  of  several  observations,  1-16 
of  arithmetical  mean,  1-16 
Product  curve,  of  permanent-magnet  materials 

2-65 

Program  amplifier,  16-09 
linos,  classes,  16-27 
service,  facilities,  17-103 
requirements,  17-101 
special  features,  17-105 
volume  range,  16-34 
Progression,  arithmetical,  1-02 

geometrical,  1-02 
Progressive  plane  waves,  5-51 

universal  winding,  3-34 
Projection  practices,  16-21 

tubes,  defined,  15-46 
Propagation  constant,  5-25 

symbol,  1-73 

wave,  Huygen's  principle,  6-76 
Properties,  of  materials,  Section  2 
Psychoses,  23-04 
Public-address  systems,  16-14 

classification,  16-14 
Pulling  figure,  4-49 
Pulsating  sphere,  13-03 
Pulse  amplifiers,  7-31,  7-55,  9-14 
coupling  time  constant,  9-14J 
i-f,  9-14 

recovery  time,  9-14 
riso  time  of  input  pulse  and,  9-14 
video  frequency,  9-15 
amplitude,    measurement  of,   by  comparison 

with  continuous  wave,  9-11 
pulac  measurements  and,  9-10 
broad  television,  20-13 
circuits,  9-13 
delay  lines,  9-28 
frequency  counters,  9-13 
frequency  dividers,  9-13 
frequency  multipliers,  9-13 
vacuum  tubes  and,  9-26 
coding,  9-02 

counter,  elementary,  9-25 
frequency  control,  8-14 
detection,  output  envelope  of  pulsed  oscillator, 

9-06 

with  background  noise,  9-05 
with  echo,  9-06 
detectors,  9-24 

double-pulse  decoder,  9-26 
duration,  9-11 
measurements,  9-10 

by  bolometer  bridge,  9-11 
by  oscilloscope,  9-10 
calorimeter  used  in,  9-12 
lamp  uyed  in,  9-12 
pulae  frequency,  9-12 
modulation,  9-05 
basic  types,  9-03 
of  an  owcillator,  9-21 
typca,  9-03 

used  in  low-speed  and  high-speed  code  trans- 
mission, 9-05 

narrowing  in  pulse  shaping  circuits,  9-15,  9-16 
power,  pulse  measurements  and,  9-10 


Pulse  selection,  9-25 

by  pulse  duration,  9-25 
coincidence  mixer  tube  and,  9-25 
shaping  circuits,  baseline  clamping,  9-17 
clamping  or  d-c  reinsertion,  9-17 
differentiation  of  pulse  in,  9-15 
narrowing  by  r-c  differentiation,  9-16 
narrowing  by  use  of  delay  line,  9-16 
narrowing  by  use  of  oscillatory  circuit,  9-16 
operating  conditions  of  tubes  in,  9-15 
pulse  narrowing  in,  9-15 
pulse  widening,  9-16 
squaring  a  sine  wave,  9-15 
systems,  communication,  9-05 
computers,  9-08 
multiplex  operation  of  several  channels  by, 

9-05 
techniques,  9-02 

electronic,  9-02 
thyratrons,  4-69 

-time  modulation,  multiplex  operation  of  sev- 
eral channels  by,  9-05 
•time  modulator,  9-23 
timing  circuits,  9-19 

blocking    oscillators,    pulse    duration    and, 

9-20 
delay  line  used  to  control  repetition  rate  of 

pulses  in,  9-19 
duration  of  pulses  and,  9-20 
transformers,  6-32,  9-27 

wave  forms,  9-27 
tubes,  9-26 
Pulsed  waves,  9-02 

comparison  of  continuous  waves  and,  9-02 
Pulses,  9-02 

core  materials  for  application,  9-28 
delayed,  9-21 

circuits,  9-21 
duration,  9-11 
equalizing,  20-13,  20-17 
frequency  spectrum  of,  9-11,  9-12 
from  diverse  locations,  9-10 
modulating,  basic  types,  9-03 

characteristics,  9-23 
phase  modulation,  9-26 
picture  transmission,  9-06 
reflection,  9-10 

repeating,  average  value  of  power,  9-11 
details,  9-12 
frequency  of,  9-12 
return,  9-10 

speed  of  information  and,  9-03 
square,  widening  of,  by  reduction  of  frequency 

bandwidth,  9-04 
synchronizing,  20-16 
time  modulation,  9-26 
timing,  distance  measured  by,  9-09 
types  of  modulation,  9-03 
use  in  measurements,  9-02 
use  in  signaling,  9-02 
Pumped  rectifier,  defined,  4-75 
Pupil  (eyes),  14-27 

Pure  coupling  in  passive  circuits,  6-07 
imaginary,  1-06 
vowels,  12-20 
Pushbutton  sets,  broad  i-f  amplifiers  used  in.  local 

osciUator  in,  7-60 
tuners,  tests  on,  11-47 
Pushing,  defined,  22-06 
PushpuU  amplifiers,  7-10 

output  transformer,  6-18 
Pylon  antenna,  6-86 
Pyralia,  2-36,  2-43 
Pyramid,  right,  mensuration,  1-18 


38 


INDEX 


Pyramidal   horn,   radiation   from    wave   guides 

formed  by,  6-63 
Pyranol,  2-52 
Pyrex,  2-43 

Pyridine,  dielectric  properties,  2-50 
Pyrotenax,  2-39 
Pyroxylin,  2-43 

0,3-48 

air  gaps,  3-49 

core  loss  and,  3-52 

for  mode  of  cavity  resonator,  7-101 

maximum,  3-49 

optimum,  3-48 

optimum  permeability,  3-49 
Q  Max.,  2-43 
Quadded  cable,  10-03 
Quadrants,  conversion  table,  1-51 
Quadratic  equation,  1-03 
Quadripoles,  equivalent,  5-13 
Quality  of  a  reactor,  symbol,  1-73 
Quantities,  complex,  1-06 

electrical,  letter  symbols  for  magnitudes,  1-72 

imaginary,  1-06 

Quantity,  electrical,  cgs  unit,  1-46 
conversion  table,  1-58 
mks  unit,  1-46 
symbol,  1-46 

of  heat,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 
Quarter,  conversion  factors,  1-54 

section,  conversion  factors,  1-48 

wave  antennas,  6-36,  6-70 
Quarts,  abbreviation,  1-72 

conversion  factors,  1-50 

liquid,  conversion  table,  1-49 
Quartz,  13-58 
'  crystal  oscillators,  7-92 

Dauphine  twinning,  13-60 

defects,  13-60 

fused,  2-43 

properties,  2-28 

physical  properties,  13-59 

power  factor  at  high  frequencies,  2-34 

principle  cuts,  13-62 

properties,  2-10 

resonators,  11-12 

use  in  oscillators,  7-92 

use  in  piezoelectric  crystals,  13-56 

useful  orientations,  13-60 
Quasi-optical  antennas,  6-63 
Quasi-steady-state  analysis  for  f-m  distortion, 

8-26 

Quiescent  point,  defined,  4-06 
Quinoline,  dielectric  properties,  2-50 
Quintal,  conversion  factors,  1-54 

Radar,  22-44 

antenna  system,  22-46 
beacons,  22-47 

marker,  22-48 

reflector,  22-48 
carrier  frequency,  22-45 
duty  cycle,  22-45 
indicator,  22-46 
installation,  22-47 
maintenance,  22-47 
maximum  range,  22-46 
minimum  range,  22-46 
monitor  for  airport  traffic  control,  22-24 
operation,  22-47 
power  supply,  22-46 
presentation,  22-47 


Radar,  principles,  22-44 
pulse  repetition  rate,  22-45 
receiver,  22-46 
resolution,  22-46 
synchronizer,  22-46 
timer,  22-46 

transmitter  pulae  width,  22-45 
transmitters,  22-46 
Radial  deflection  tubes,  15-47 
Radians,  conversion  table,  1-51 
Radiated  powers,  5-52 
Radiation,  5-49 
acoustic,  13-03 
efficiency  of  antennas,  6-68 
electromagnetic,  5-49 
in  free  space,  10-29 
of  microwave  antennas,  6-77 
resistance  of  antennas,  6-68 
secondary,  23-15 
soft,  of  x-ray,  4-82 
thermal,  15-29 
Radiator,  elementary,  5-52 
Radio  antennas,  6-62 

flat-top  of,  6-62 

general  function  and  description,  6-62 

lead-in,  6-62 

loop,  6-62 
broadcast,  16-25 
frequencies,  11-14 

amplifiers,  7-22,  7-24 
markers,  22-09 
-phare,  22-05 
police,  16-35 

equipment,  16-37 

frequency  allocation,  16-36 

power  and  range,  16-37 
propagation,  10-29 
range,  defined,  22-05 

low-frequency,  22-06 

visual  two-course,  22-08 
receivers,   7-115;   see   also   Receiver   circuits; 
Receivers 

antenna  coupling  circuit,  7-115 

functions,  7-115 

gain  and  selectivity  sources,  7-58 

high-inductance  antenna  coupling  of,  7-115 
spectrum,  general  view,  10-29 
stations,  range  of,  10-47 
studios,  12-02 

noise  levels  acceptable  in,  12-58 
telegraph,  control  channels,  18-60 
bandwidths,  18-61 
recorder  drive,  18-61 
tone  keyers,  18-60 
tone  signal  converters,  18-60 

fading,  reduction  of,  18-57 

frequencies,  18-56 

frequency  shift  keying,  18-58 

interference,  18-58 

receiver  sites,  18-56 

traffic  office,  equipment,  18-58 

transmitter  sites,  18-56 
telephone  broadcasting,  16-25 

fidelity  requirements  of  system,  16-33 

frequency  allocation,  16-30 

program  distribution  systems,  16-27 

program  lines,  16-27 

station  service,  16-32 

transmitter  plant,  16-28 
telephone  systems,  17-54 

coastal  harbor  and  inland  waterways,  17-59 

highway  mobile,  17-59 

installation,  17-66 

railway  mobile,  17-60 


INDEX 


39 


Radio  telephone  systems,  rural  subscriber,  17-60 
special  emergency,  17-61 
two-way  operation,  17-62 
interconnecting  with  two-wire  extensions, 

17-63 

with  four-wire  terminals,  17-62 
telephone  transmission,  17-61 
operational  methods,  17-61 
privacy,  17-62 
requirements ,  17-65 

single  sideband,  suppressed  carrier,  17-61 
spread  sidebands  and  carrier,  17-61 
two  sidebands  and  carrier,  17-61 
two  sidebands,  suppressed  carrier,  17-61 
transmission,  national  and  international  regu- 
lations, 7-120 
transmitters,  7-129 
a-rn,  7-120 

audio  amplifiers,  7-134 
circuit  Q  and,  7-130 
frequency  control,  7-129 
harmonic  amplifiers  and,  7-130 
high-power,  7-136 

audio  equipment,  7-136 
incidental  phase  modulation,  7-136 
installation,  7-134 
intermodiate-r-f  amplifiers,  7-129 
interstage  coupling  circuits,  7-129 
multiple  resonances  and,  7-129 
parasitics  and,  7-129 
low-power,  7-136 
modulation    characteristics    measurements, 

7-185 

negative  feedback  and,  7-133 
oscillator  power,  7-129 
power  amplifiers  and,  7-131 
rectifier  and  power  equipment,  7-137 
scope,  7-129 

fltibfltation  for  power,  7-137 
unmodulated  intermediate  amplifiers,  7-130 
waverneter,  11-12 
waves,  pulsed,  9-02 
Radiohm,  properties,  2-08 
Radiometer,  15-04 
Radiosondes,  22-13 
Radiothermy,  23-04 
Range,  auditory,  9-06 
muwo,  12-30 
of  radio  station,  1 0-47 
Ranges,  standard  r-f,  1-80 
Raster,  20-03 

Rate  of  doing  work,  conversion  table,  1-58 
Ratio,  aopect,  204)3,  20-20 
f-m  detection,  distortion  and,  8-29 
logarithmic  voltage,  1-38 
of    electrostatic     to    electromagnetic    unite, 

1-79 

of  maas  of  H  to  mass  of  electron,  1-79 
physical  constants  and,  1-79 
-typ©  frequency  detectors,  8-21 
Rationalized  mke  units,  1-45 
Ray  acoustics  of  rooms,  12-41 
Rayo,  properties,  2-08 
Rayon,  2-43 
R-c  coupled  amplifiers,  7-04,  7-90 

tuned  oscillators,  7-90 
RCA  broadside  arrays  antennas,  6-65 
Reactance,  oapacitative  r.,  symbol,  1-73 
coupling,  6-07,  6-08,  6-09 
function  of  networks,  6-35 
inductive  r.,  symbol,  1-73 
minimum,  5-08 
mutual  r.,  symbol,  1-73 
self  r.,  symbol,  1-73 


Reactance,  symbol,  1-73 

tubes,  frequency  modulation  by,  8-12 
Reactive  component  of  input  admittance,  4-18 
factor,  symbol,  1-73 
kilovolt-ampere,  abbreviation,  1-72 
power,  cgs  unit,  1-46 

mks  unit,  1-46 

symbol,  1-46 

volt-ampere,  abbreviation,  1-72 
Reactors,  audio  frequency  plate,  3-48 
rectifier-filter,  3-48 
saturable,  3-48 

Rear  feed  for  reflectors  of  antennas,  6-86 
Receiver  circuits,  compensated  volume  control, 
7-126 

distribution  of  amplitude,  7-128 

distribution  of  envelope  amplitude  vs.  time, 
7-128 

effect  of  audio  transformer  on  fidelity,  7-126 

random  noise,  nature  of,  7-128 

tone  control,  7-126 
power  supply,  7-106 

typical,  7-107 

Receivers,  7-115;  see  also  Radio  receivers;  Re- 
ceiver circuits 

a-c  operated,  filament  power  for,  7-106 
aircraft,  22-21 
all-wave,  7-124 
audio  output,  7-124,  7-125 
automatic  direction  finder,  22-11 
automatic  volume  control,  advantage  of,  7-125 
crystal  detector,  7-117 
fidelity  characteristics,  7-125 
f-m,  see  F-m  receivers 
f-m — a-m,  8-17 

circuit  diagram,  8-18 
Johnson  noise,  7-127 
neutralized,  7-119 
noise  factor,  7-127 
noise  suppression,  7-126 
noise  suppressor  circuit,  7-127 
one-tube  superregenerative,  7-118 
picture,  9-07 
radio,  see  Radio  receivers 
random  noise,  7-127 

receiving  tubes  in,  voltage  supply  for  plate  cir- 
cuits of,  7-106 
reception   of   continuous   wave   code   signals, 

7-124 

reduction  of  effect  of  fading  signals,  7-125 
regenerative,  7-117 

objection  to,  7-117 

oscillation,  7-117 
regenerative  detector,  7-124 

circuit,  7-124 

resistance-coupled  audio  amplifier,  7-125 
shot  effect  in  vacuum  tubes,  7-127 
six-tube  superheterodyne,  7-120 
superheterodyne,  combined  first  detector  and 
oscillator,  7-119 

frequency  converter,  7-119 

preselector,  7-119 

superregeneration  of  the  blocking  type  in,  7-117 
telephone,  see  also  Telephone  receivers 

graphical  symbol,  1-76 

simple,  5-65 
television,  20-46 

thermal  agitation,  in  circuit  resistances,  7-127 
tuned  r-f,  7-118 

neutralization  use  to  eliminate  oscillation, 
7-119 

regeneration  in  multistage  amplifiers,  7-118 
tuning  indicators,  7-124 
types,  7-117 


40 


INDEX 


Receiving  tube  classification  chart,  4-28 

Reciprocal  impedances,  5-09 

Reciprocity  theorem,  for  linear  networks,  5-12 

of  Lord  Rayleigh,  10-29 
Recording  practices,  16-19 

studios,  noise  levels  acceptable  in,  12-58 
polycylindrical  sound  diffusers,  12-71 
sound  insulation,  12-57 
wave  analyzer,  11-65 

Recordings,  see  Magnetic  recordings;  Mechanical 
recordings ;  Photographic  sound  recordings 
Recovery  time  in  pulse  amplifier,  9-14 
Rectangular  hyperbola,  equation,  1-05 
wave  guides,  10-11 

bends,  twists,  and  angles,  10-21,  10-22 
Rectification  diagrams,  7-79 
Rectifier  circuits,  7-110 

class  B  modulator,  7-113 
control  systems,  7-114 
double  output,  7-110 
double-stage  filter,  7-112 
filter  chokes,  7-113 
high-voltage  transformers,  7-110 
rectifier  control  systems,  7-114 
rectifier-tube  operation,  7-113 
single-stage  filter,  7-112 
telephone,  linear  amplifier,  7-112,  7-113 
tube,  1-76,  4-28 
tube  heater  delay,  7-114 
tube-failure  prediction,  7-114 
wave  form,  7-107 
pumped,  defined,  4-75 
tubes,  mercury-vapor,  7-108 
output  characteristics,  7-108 
supply  filter,  7-108 
Rectifiers,  3-25 
application,  21-03 
circuit,  21-03 
classification,  4-03 
contact,  11-17 
controlled,  21-09 
double  output,  7-110 
electronic,  21-02 
gas-filled,  4-03 
high  vacuum,  4-03 
ignitron,  21-10 
mercury-arc,  4-77 
non-controlled,  21-08 
transformers  used  in,  7-110 
types  used  in  transmitters,  7-109 
vacuum  tube,  7-106 

Recurrent  networks,  see  Networks,  recurrent 
Red  cadmium  line,  wave  length,  1-79 
Redray,  properties,  2-08 
Reference  level,  1-41 
noise,  11-37 

pressure  of  sound,  12-05 
volume,  16-10 
Reflected  component,  5-23 

waves,  symmetrical  networks,  5-23 

uniform  lines,  5-25 
Reflecting  field,  4-54 
Reflection,  14-02 

at  spherical  surfaces,  14-05 
coefficient  plane,  4-56 
diffuse,  14-04 
factor,  5-15 
law,  14-03 
losses,  5-16 

of  pulse  in  distance  measurement,  9-10 
Reflectivity,  acoustic,  12-40 
Reflectors  of  antennas,  6-78 
Reflex  klystrons,  4-54 

tubes,  external  cavity  type,  4-57 


Reflex  tubes,  internal  cavity  type,  4-57 
Refraction,  14-02 
at  spherical  surfaces,  14-05 
dispersion,  14-05 
index,  5-53 
law,  14-02 
sound,  12-60 
Refrax,  properties,  2-10 
Regeneration  in  multistage  amplifiers  of  tuned 

r-f  receivers,  7-118 
Regenerative  receivers,  7-117 
objection  to,  7-117 
oscillation,  7-117 

Register  ton,  conversion  factors,  1-50 
Regulators,  voltage,  4-08 
Relative  capacity,  symbol,  1-72 
permeability,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 
Relaxation  circuits,  9-17 
blocking  oscillator,  9-18 
flip-flop  circuit  stable  in  either  condition, 

9-17 

free-running  multivibrator,  9-18 
triggered  multivibrator,  9-18 
oscillation,  7-84 

Reluctance,  magnetic,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 
symbol,  1-73 
Reluctivity,  symbol,  1-73 
Remalloy,  2-66,  2-67 
Remote  cutoff  tube,  4-25 
Repeater,  7-13 
Repeller,  4-54 
Repetition  test,  12-28 
Reproducing  of  sounds,  13-37 
Reproduction  systems,  16-02 
Residual  induction,  defined,  2-60 
Resimene,  2-39,  2-43 
Resinox,  2-43 
Resins,  acrylic,  2-34 
alkyd,  2-34 
allyl,  2-34 

Resistance,  acoustic,  13-02 
arc,  2-24 

capacitance  coupling,  7-04 
-coupled  audio  amplifier  of  receivers,  7-125 
coupling,  6-07,  6-08,  6-09 
electrical,  cgs  unit,  1-46 
conversion  table,  1-61 
mks  unit,  1-46 
symbol,  1-46 
electrode,  4-11 
equivalent  negative,  7-28 
equivalent-noise,  4-21 
grid,  4-11 
insulation,  2-23 
lamps,  3-22 
loads,  5-42 
measurement,  11-23 

measurements  of,  audio-frequency  transform- 
ers, 6-25 
minimum,  5-08 
mutual  r.,  symbol,  1-73 
plate,  4-11 

radiation,  6-68,  13-02 
self  r.,  symbol,  1-73 

stabilization  in  tuned  r-f  receivers,  7-118 
standard,  11-18 
decade,  11-20 
symbol,  1-73 

-temperature  coefficient,  symbol,  1-73 
Resistivity,  defined,  2-02 


INDEX 


41 


Resistivity,  earth,  10-32 

electrical,  cgs  unit,  1-46 
conversion  table,  1-61 
mks  unit,  1-46 
symbol,  1-46 

surface,  2-23 

symbol,  1-73 

thermal,  conversion  table,  1-65 

units,  defined,  2-02 

volume,  2-23 
Resistoflex,  2-43 
Resistors,  3-02 

adjustable  or  variable,  graphical  symbol,  1-76 

classifications,  3-04 

dose-tolerance,  3-06 

color  code,  3-12 

composition  carbon,  3-11 

defined,  3-02 

deposited-carbon,  3-15 

energy  dissipation,  3-03 

fixed  resistance,  3-11 

flat-type,  3-09 

flexible,  3-09 

frequency  characteristic,  3-03,  3-14 

graphical  symbol,  1-76 

in  printed  circuits,  3-22 

low-power,  3-07 

metal  film,  3-17 

non-ohmic,  3-23 

pad-type,  3-09 

physical  and  electrical  considerations,  3-02 

plug-in  typo,  3-09 

power  ratings,  3-03 

power-type,  3-05 
layer  windings,  3-07 
non-inductive  windings,  3-07 

precision-type,  3-07 

resistance  tolerance,  3-03 

resistance  value,  3-02 

special-purpose,  3-21 

specifications,  3-05 

spool  type,  3-09 

stability  with  age,  temperature,  and  humidity, 
3-04   , 

temperature  rise,  3-03 

tests,  3-05 

thermal,  typical  characteristics,  11-77 

winding  typos,  3-09,  11-19 

wire-wound,  3-05 

Resolution,  television,  20-02,  20-06,  20-66 
Resonance,  5-30,  6-02 

analyzers,  11-58 
effect  of,  12-33 

frequency  formulas,  11-03 

frequency  motor,  11-11 

measurements,  11-27 

room,  12-45 
Resonant  antennas,  6-81 

circuits,  6-02 

magnetron,  graphical  symbol,  1-78 

-V  antennas,  6-83 
Resonators,  4-42 

cavity,  tfae  Cavity  resonators 

quartz,  11-12 

unsealed,  humidity  changes,  11-88 
Respiration,  artificial,  23-18 
Respiratory  muscles,  speech  and,  12-19 
Restaurants,  noise  levels  acceptable  in,  12-58 
Restrictions  for  physical  readability,  5-11 
Resuscitation,  23-18 
Retina,  of  oyoa,  14-25 
Retrace,  television,  20-06 
Return  of  pulse  in  distance  measurement,  9-10 
Reverberation,  12-40 


Reverberation,  at  different  frequencies,  12-47 
combined  effects  of  loudness  and,  12-74 
effect  on  hearing  of  speech,  12-73 
equations,  12-43 
formula,  modification  of,  12-43 
free  decay  of  modes  of  vibration  and,  12-45 
in  auditoriums,  12-73 
in  coupled  spaces,  12-46 
measurement,  12-48 
percentage  articulation  for  various  size  rooms 

and  different  times  of,  12-74 
room,  effect  on  articulation,  12-38 
time,  12-40,  12-43,  12-44,  12-45,  16-03 

optimal,  12-75,  16-03,  16-11 
Reversible  permeability,  defined,  2-60 
Revolutions  (circumferences),  conversion  table, 

1-51 
Revolutions  per  minute,  abbreviation,  1-72 

per  second,  abbreviation,  1-72 
R-f  amplifiers,  7-22 

circuits,  effect  of,  on  fidelity  curve  of  receivers, 

7-125 

gain  and  bandwidth,  11-53 
signal,  20-17 
Rheostats,  3-02,  3-17 
defined,  3-02 

physical  and  electrical  considerations,  3-02 
power-type,  3-19 
wire-wound,  3-18 
Rheotan,  properties,  2-08 
Rhodium,  properties,  2-08 
Rhombic  antennas,  6-64,  6-83 
Ribbon  microphone,  13-23 

speaker,  13-11 

Rice  system  for  neutralization,  7-29 
Richardson's  temperature  law,  4-09 
Rickets,  23-08 
Rieke  diagram,  4-49,  4-56 
Right  circular  cone,  mensuration,  1-18 
circular  cylinder,  mensuration,  1-18 
cylinders,  modes  in,  mode-shape  factor,  7-101 
pyramid,  mensuration,  1-18 
rectangular  cylinder   cavity  resonator,    7-96, 

7-99,  7-101 
Rigid  partition,  12-65 
Ripple  frequency,  7-109 
Rise  time  of  input  pulse  in  amplifiers,  9-14 
Rising  sun  anode  block,  4-42 
RMA  color  code,  3-12 

preferred  numbers,  3-12 
Rochelle  salt,  13-56 
properties,  13-65 
use  in  oscillators,  7-92 
use  in  piezoelectric  crystals,  13-56 
useful  cuts,  13-67 
Rocket  antennas,  6-85 
Rod,  conversion  factors,  1-47 

square,  conversion  factors,  1-48 
Roebling  gage,  1-69 
Roentgen  therapy,  apparatus,  23-12 
van  de  Graaff  generator,  23-12 
general  technical  requirements,  23-12 
purpose,  23-12 
technique,  23-13 

Roentgenography,  apparatus,  23-15 
general  technical  requirements,  23-14 
industrial,  tubes,  4-89 
medical,  tubes,  4-86 
purpose,  23-14 
technique,  23-14 

Roentgenoscopy,  apparatus,  23-15 
general  technical  requirements,  23-14 
purpose,  23-14 
technique,  23-14 


42 


INDEX 


Roentgenoscopy  tubes,  4-86 
Roods,  conversion  factors,  1-48 
Rooms,  acoustic  properties,  12-39 

banking,  noise  level  acceptable  in,  12-58 

conference,  noise  levels  acceptable  in,  12-58 

growth  and  decay  of  sound,  12-42 

lecture,  12-42 

live,  reverberation  equations  and,  12-43 

music,  see  Music  rooms 

normal  frequencies  for  acoustics,  12-45 

percentage  articulation,  12-69 

resonance,  12-45 

resonant  frequencies  for  acoustics,  12-45 

speech  power  of  speakers  in  various  sizes,  12-73 
Root-mean-square,  1-74 

abbreviation,  1-72 
Ropes,  conversion  factors,  1-47 
Rose's  metal,  properties,  2-08 
Rosin,  2-43 

oil,  dielectric  properties,  2-50 

properties,  2-28 

Rotary  joints  of  wave  guides,  10-26 
Rotational  frequency,  symbol,  1-73 
Rotative  operators,  symbols,  1-73 
R-63  alloy,  properties,  2-08 
Rubber,  cyclicized,  2-43 

hard,  2-43 
power  factor  at  high  frequencies,  2-34 

properties,  2-28,  2-30 

sponge,  compliance  and  resistance  data  in  insu- 
lation, of  vibration,  12-62 

synthetic,  properties,  2-30 

vulcanized,  2-44 
Rubidium,  properties,  2-08 
Rug,  12-54 
Rutile,  2-45 

Sabin  of  acoustic  properties  of  rooms,  12-40 
Saflex,  2-45T  2-47 
Saran,  2-45,  2-47 

properties,  2-30 
Saturable  reactors,  3-48 
Saturated  sleeving,  2-45 
Saturation  induction,  defined,  2-60 

permeameter,  2-74 
Saxophone,  12-25,  12-30 
Scala  media  of  ear,  12-03 

tympani  of  ear,  12-03 

vesitibuli  of  ear,  12-03 
Scales,  temperature,  1-43 
conversion  table,  1-64 
Scaling,  magnetrons,  4-45 
Scalloping,  denned,  22-06 
Scanning,  20-03 

circuits,  for  cathode-ray  devices,  20-08 
for  electrostatic  deflection,  20-08 

horizontal  resolution,  20-04 

interlaced,  20-05 

keystone  correction,  20-11 

magnetic,  circuits,  20-10 
output  amplifiers,  20-61 

mechanical,  20-07 

method   of   microwaves   in   direction  finding, 
6-87,  6-88 

vertical  resolution,  20-04 
Schematic  eye,  14-28 
Schering  bridge,  11-26 
Schools,  address  systems,  16-18 

sound  insulation,  12-57 
Scophony,  20-08 
Scotch  tape,  2-45 
Scotophors,  characteristics,  15-37 

defined,  15-32 

mechanisms,  15-36 


Screen-grid,  defined,  4-04 
lead  inductance,  4-17 
tubes,  4-03,  6-11 

Screens,  cathode-ray-tube,  15-37 
intensifying,  23-15 

luminescent,  see  Luminescent  screens 
Scruples,  conversion  factors,  1-55 
Sealed  tube,  defined,  4-75 
Secant,  1-07 

abbreviation,  1-72 

hyperbolic,  1-10,  1-26 
Sech,  WO,  1-26 
Second  (time),  abbreviation,  1-72 

standard,  1-43 

Second  (angular  measure),  abbreviation,  1-72 
Second  harmonic  distortion,  4-24 

in  f-m  systems,  8-28 
Secondary  daytime  coverage,  16-32 

emission,  defined,  4-05 

night-time  coverage,  16-32 

radiation,  23-15 

service,  16-32 

Seconds  (angle),  conversion  table,  1-51 
Seconds  (time),  conversion  table,  1-51 
Section  of  land,  conversion  factors,  1-48 
Sectionally  wound  coils,  use  in  i-f  amplifiers,  7-58 
Selectance,  7-115 
Selective  fading,  multipath  transmission  and, 

8-29 

Selectivity,  determination,  in  superheterodyne  re- 
ceivers, 7-56 

in  coupled  circuits,  6-10 
Selectron,  2-39 
Self-admittance,  5-06 
Self-excited  oscillator,  4-49 
Self-impedances,  6-07 
Self-inductance,  symbol,  1-73 
Self-oscillation,  conditions,  7-84 
Self-reactances,  in  coupled  circuits,  6-07 
Self-resistances,  in  coupled  circuits,  6-07 
Semiabsolute  system,  1-45 
Semi-butterfly  circuit,  3-57 
Semicircular  canal  of  ear,  12-02 

plate  forms,  3-57 
Semi-remote  cutoff  tube,  4-26 
Semi-tone,  11-09 
Sendust,  2-62 
Sensation  area,  auditory,  12-08 

unit,  1-38 

Sensitivity,   determination,    in   superheterodyne 
receivers,  7-56 

ear,  12-08 

Sequential  television,  20-65 
Series  expansion,  for  plate  current  of  a  triode, 
5-41 

mathematical,  1-17 

plate  circuit,  5-41 

power,  5-38 

resonant  circuits,  6-02 

Taylor's,  1-17,  5-38 

trigonometric,  5-39 

-type  networks,  6-35,  6-36 
Service  broadcast  range,  10-47,  16-32 
Shape  of  room,  effect  of,  12-69 
Sharpness,  20-02 

roentgenographic,  23-14 
Sheet  metal  gages,  1-66 
Shellac,  2-45 

properties,  2-30 

Shield,  within  envelope,  graphical  symbol,  1-78 
Shielded  loop  antennas,  6-87 
Shielding  cans,  input  and  interstage  transformers 

and,  6-21 
large,  use  in  i-f  amplifiers,  7-58 


INDEX 


43 


Shielding,  problems,  11-101 

Shin,  1-10, 1-26 

Shipping  measure,  conversion  factors  1-50 

ton,  1-50 
Shock,  acoustic,  10-92 

electric,  23-17 
Shock  therapy,  23-02,  23-04 
Shockproof  x-ray  diffraction  tube,  4-89 
Short  ton,  conversion  factors,  1-54 
wave,  10-40 
-wave  antennas,  locations,  6-75 

sets,  7-60 
Shot  effect,  4-20 
Shunt-fed  antenna,  6-81 

neutralization  employed  in  power  amplifiers 

7-132 

peaked  amplifiers,  7-33 
circuit,  7-33 
inductance  in  terms  of  frequency  and  plate 

load,  7-34 

plate  load  in  terms  of  frequency,  7-33 
Side  responses,  reduction  of,  in  f-m  distortion, 

8-29 

Sideband  analysis-synthesis  method,  for  evaluat- 
ing f-m  distortion,  8-27 
Sidebands  of  carrier  frequency,  in  process  of 

modulation,  7-70 

Side-tuned  circuits  as  frequency  detectors,  8-21 
Siemens,  mho,  1-46 
Signal  generators,  11-30,  11-89 
amplitude  modulation,  11-93 
frequency  modulation,  11-94 
noise  measurement,  11-44 
pulse  modulation,  11-95 
-to-intorforence  ratio  in  f-m  systems,  8-30 
tranaadmittance,  4-55 
Signaling,  use  of  pulses,  9-02 
Signals,  Naval  Observatory  time,  18-40 
selection  in  frequency  or  time,  9-02 
telegraph,  18-05 
Silaneal,  2-45 
Silastic,  2-45 

properties,  2-30 
Silchrome,  properties,  2-08 
Silfrax  B,  properties,  2-11 
Silica,  fused,  2-45 
Silicon,  2-45,  2-53 
-iron  alloys,  2-61 
powder,  properties,  2-11 
properties,  2-08,  2-11 
Silicone  fluid,  properties,  2-30 
dielectric  properties,  2-50 
glass  laminate,  properties,  2-30 
rubber,  properties,  2-30 
sealing  compound,  properties,  2-30 
varnish,  properties,  2-30 
varnished  glass  cloth,  properties,  2-30 
Sillc,  2-40 
Silmanal,  2-68 

Silver  chloride,  properties,  2-11 
electrochemical  equivalent,  1-79 
properties,  2-08,  2-11 
Simple  raster,  20-03 

aeries  circuit,  variation  of  current  with  fre- 
quency, 6-03 

variation  of  voltage  components  with  fre- 
quency, 6-03 
Simultaneous  equations,  1-03 

linear,  1-04 
Sine,  1-07 

abbreviation,  1-72 
anti-hyperbolic,  1-10 
hyperbolic,  1-10 
abbreviation,  1-71 


Sine,  hyperbolic,  inverse,  1-10 
Single-anode  tube,  defined,  4-75 
-frequency  modulation,  8-02 
-frequency  tone  (sounds),  12-14 
-mesh  circuits,  5-02,   5-03,  see  oho  Circuits, 

single-mesh 
Sinh,  1-10,  1-26 
Sink,  frequency,  4-56 

margin,  4-56 
Sinusoid,  equation,  1-05 
Sisal  hemp,  2-46 
Skin  effect,  3-04,  11-18 

eruptions,  23-08 
Skip  distance,  10-41 

transmission,  f-m  distortion  and,  8-29 
Sky  wave,  see  Wave  propagation,  sky  wave 
Slate,  properties,  2-30 
Slide-wire  rheostat,  11-19 
Slip,  symbol,  1-73 
Slope  filters,  f-m  receivers  and,  8-19 

time,  5-35 

Small  deviation  ratio,  in  f-m  distortion,  8-27 
Smith  chart,  4-56 
Snell's  law,  14-02 
Sodium  chloride,  properties,  2-11 

properties,  2-08 
Soft  x-rays,  4-82 

Soil,  effect  of,  on  directivity  of  antennas,  6-74 
Solar  day,  1-43 

disturbances,  10-46 
Solid  angle,  conversion  table,  1-51 

minimum  resolvable,  20-02 
anode  magnetron,  4-40 
Solids,  dielectric  absorption,  2-23 

dielectric  strength,  2-23 
Sonne,  22-31 
Sound  absorption,  coefficients,  12-50 

of  audience,  individual  persons  and  other 

objects,  12-55 

of  hair  felt  in  different  laboratories,  12-56 
of  thicknesses  of  hair  felt,  12-56 
practical  considerations  of  materials,  12-57 
-absorptive  coefficients  of  materials,  12-48 
carrier,  20-20 
chamber,  13-07 
film,  distortion,  12-35 
foci,  12-70 
insulation,  12-57 

by  rigid  partitions,  12-63 
ceiling  isolator  used  in,  12-60 
connectors  used  in,  12-60 
considerations  in  selection  of  building  mate- 
rials and  types  of  structure  for,  12-67 
flexible  cushions  used  in,  12-60,  12-61 
floor  chair  used  in,  12-60 
method  for  insulation  of  vibration,  12-61 
principles,  12-60 
supports  used  in,  12-60,  12-61 
transmission  coefficient  for  masonry  or  con- 
crete, 12-63 

types  of  structures  recommended,  12-68 
intensity,  relation  between  nerve  discharge  rate 

for  a  1050-cycle  tone  and,  12-04 
sensitivity  of  ear  and,  12-05 
recording,  16-19 
-reinforcing  systems,  16-14 
indoor,  16-15 
outdoor,  16-15 

-reproduction  systems,  16-02 
studios,  16-02 
noise  levels,  16-04 
reverberation  time,  16-03 
sound  decay  rate,  16-04 
systems,  audio  amplifiers,  16-06 


44 


INDEX 


Sound  systems,  audio  facilities,  16-02 

control  equipment,  16-06 
transmission,  coefficients,  12-64 

reduction  factors,  12-65,  12-66 
velocity  of,  13-02 
Soundhead,  16-22    i 
Sounds,  see  also  Acoustics 

absorption  in  air  of  rooms  and,  12-42 
articulation,  12-27 
auditory  perspective,  12-39 
common  speech,  characteristics,  12-20 
growth  and  decay  of,  in  rooms,  12-42 
impact,  conduction  of,  12-60 
individual  sound  articulation,  12-27 
iso-masking  intensity,  12-1 1 
judgments  of  half  pitch,  12-16 
localization,  12-18,  16-02 
loudness,  12-11,  12-14,  16-13 

auditory  magnitude,  12-11 

computation   for,   with    continuous   energy 
spectrums,  12-13 

defined,  12-11 

neural  pulses  and,  12-11 

sone  density,  12-13 

sone  of  auditory  magnitude,  12-11 
loudness  of  levels,  12-11 
loudness  vs.  loudness  level,  12-12 
loudness-level  contours,  12-15 
magnitude  of  subjective  harmonics  and,  12-15 
masking  contours  for  steady  noise,  12-11 
masking  effects  of,  12-11 
masking  spectrums,  12-11 

of  single-frequency  tone,  12-15 
perspective,  16-02 
power,  16-18 
pressure,  16-04 
projection,  16-21 
recording,  16-19 

reflection  from  domed  ceiling,  12-70 
relative  occurrence  in  telephone  conversation, 

12-20,  12-21 
single-frequency  tone,  12-14 

stimulation  patterns,  12-15 
sound  articulation,  12-27 
spectrums  of,  in  speech,  12-20 
speech,  characteristics,  12-20 

damping  constants,  12-20 

low-frequency  modulation,  12-20 
steady,  pitch,  12-16 
syllable  articulation,  12-27 
transmission    of,    through    ventilating    ducts, 

12-61 

vowel  articulation,  12-27 
vowel-like,  of  speech,  12-20 
Space  capacitivity,  cgs  unit,  1-46 

mlcs  unit,  1-46 

symbol,  1-46 
charge,  4-02 
-charge  cloud,  4-41 
-charge  grid,  defined,  4-04 
permeability,  1-44 

cgs  unit,  1-46 

mks  unit,  1-46 

symbol,  1-46 
sense,  14-39 

direction  localization,  14-39 
transmission,  10-29 
Span,  conversion  factors,  1-47 
Spark  gap,  plain,  graphical  symbol,  1-76 
oscillator,  7-94 
quenched,  symbol,  1-76 
rotary,  1-76 
Sparkover,  2-54 

voltages  for  sphere  gaps,  2-55 


Spauldite,  4-46 
Speakers,  condenser,  13-16 
direct  radiator,  13-11 

enclosures,  13-13 
dynamic,  13-11 
horn-type  moving-coil,  13-14 
loud,  13-07 
magnetic,  13-11 

balanced-armature,  13-15 
magnetic-armature,  13-15 

bipolar,  13-16 
moving-conductor,  13-11 
pneumatic,  13-17 
Special-purpose  resistors,  3-21 
Specific  gravity,  abbreviation,  1-72 
heat,  abbreviation,  1-72 

conversion  table,  1-64 
inductive  capacity,  electrical,  cgs  unit,  1-46 
mks  unit,  1-46 
symbol,  1-46 
Spectrographs,  11-65 

sound,  11-68 

Spectrum  analyzers,  5-28 
electrical,  11-14 
noise,  11-64 

of  sounds  in  speech,  12-20 
Speech,  see  also  Acoustics;  Sounds;  Speech  power 
analysis,  12-23 
and  music,  12-19 
articulation  test,  12-27 
audible  effects  of  phase  distortion,  12-37 
audible  frequency  range,  12-30 
average  power  of  speakers  in  various  sizes  of 

rooms,  12-73 

characteristics  of  sounds,  12-20 
conversational,  speech  pressure  vs.  frequency 

range,  12-22 

description  of  organs,  12-19 
effects  of  distortion,  12-29 
hearing  of,  effect  of  noise,  12-72 

effect  of  reverberation,  12-73 

in  auditoriums,  12-69 
inflection,  12-20 
input  equipment,  12-02 
larynx  and,  12-19 

levels  of,  average,  for  speakers  in  various  sizes 
of  rooms,  12-73 

ratio  of  noise  to,  12-72 
lungs  and,  12-19 
mechanism,  12-19 

breath  steam  as,  12-19 

friction  modulation,  12-19 

modulation,  12-19 

vocal-cord  modulation,  12-19 
mouth  cavities  and,  12-19 
nose  cavities  and,  12-19 
optimal  reverberation  times  for,  12-75 

at  different  frequencies,  12-75 
pharynx  and  characteristics  of  sounds,  12-20 
power,  12-22 

frequency  distribution,  12-24 

phonetic  powers  of  1  watt,  12-24 

pressure  measurement  of  mouth  as  radiator, 

12-22 

•    powers,  data  on,  12-23 
pressures,  11-60 
production,  12-19 
respiratory  muscles  and,  12-19 
spectrum,  12-23 
stress,  12-20 
tests,  12-27,  12-28 

laboratory,  12-27 
throat  cavities  and,  12-19 
trachea  and,  12-19 


INDEX 


45 


Speech,  transmission,  12-27 
vibrato,  12-20 
visible,  12-21 

cavity  modulation,  12-21 
fill,  12-21 

frietional  modulation,  12-21 
patterns,  12-22 
phonetic  printing  and,  12-21 
plosive  release,  12-21 
start-stop  modulation,  12-21 
stop  gap,  12-21 
vocal  resonance  bars,  12-21 
voice  bar  modulation,  12-21 
visual,  12-21 

vocal  cords  in,  glottis,  12-19 
vooal  resonances,  12-20 
vocoder  used  in,  12-21 
voder  used  in  artificial,  12-21 
vowels,  12-20 
wave  forms,  12-22 
average,  12-22 
instantaneous ,  12-22 
peak,  12-22 
types  denned,  12-22 
windpipe  and,  12-19 
Speed  of  information,  5-28 

frequency  bandwidths  required  for,  9-04 
pulses  and,  9-03 
of  rotation,  1-73 
Spheres,  conversion  table,  1-51 

mensuration,  1-18 

Spherical  candlepower,  abbreviation,  1-72 
optics  of  antennas,  6-77 
right  angles,  conversion  table,  1-51 
Sponge  rubber,  12-62 
Spot,  scanning,  20-04 
Square,  abbreviation,  1-72 
centimeters,  abbreviation,  1-72 

conversion  table,  1-48 
conversion  factors,  1-48 
cross  bracket,  5-43 
factors,  conversion  factors,  1-48 
foot,  abbreviation,  1-72 
conversion  table,  1-48 
pounds  per,  abbreviation,  1-72 
inch,  abbreviation,  1-72 
conversion  table,  1-48 
pounds  per,  abbreviation,  1-72 
kilometer,  abbreviation,  1-72 
kilometers,  conversion  table,  1-48 
-law  detection,  7-76,  7-78 

defined,  7-76 
-law  detectors,  detection  of  carriers  and,  7-76 

single  sideband  signals  and,  7-7ft 
motor,  abbreviation,  1-72 
conversion  table,  1-48 
micron,  abbreviation,  1-72 
mile,  conversion  table,  1-48 
millimeter,  conversion  table,  1-48 
myriameter,  conversion  factors,  1-48 
pulses,  widening  of,  by  reduction  of  frequency 

bandwidth,  9-04 
rod,  conversion  factors,  1-48 
root  of  mean  square,  abbreviation,  1-72 
wave  testing,  11-39 
yard,  conversion  table,  1-48 
Squares,  theory  of  least,  1-15 
Stability,  frequency,  7-92 
Stabilization,  receiver,  7-118 
Stacking  of  arrays,  antennas  and,  6-73 
Stadium,  address  system,  16-15 
Stage,  amplifier,  7-04 
sound,  12-02 


Staggered  tuning  of  coupled  resonant  circuits, 

6-10 
Stagger-tuned  single-tuned  circuits  of  amplifiers, 

7-64 

Stainless  type  304,  properties,  2-08 
Standard  broadcasting,  16-30 
coverage,  16-32 

capacitance,  11-20 

time,  11-04 

Wire  Gage,  1-69 
Standards,  abbreviation,  1-72 

of  fundamental  units,  1-42 

television,  20-20 
Standing-wave  detectors,  11-72 

waves,  uniform  lines,  5-25 
Starting  voltage,  4-08 

Start-stop  modulation  in  speech,  12-19,  12-20 
Stat,  1-43 

Statamperes,  conversion  factors,  1-59 
Statcoulombs,  1-43 

conversion  factors,  1-58,  1-59 
Statfarads,  1-43 

conversion  factors,  1-62 
Stathenrys,  conversion  factors,  1-63 
Static-characteristic  measuring  circuit,  4-08 

plate  resistance,  5-42 
Statohms,  conversion  factors,  1-61 
Statute  miles,  conversion  factors,  1-47 
Statvolts,  conversion  factors,  1-60 
Steady-state  velocity,  5-36 
Steatite,  2-46 

ceramics,  properties,  2-30 

power  factor  at  high  frequencies,  2-34 
Steel  wire,  copper-clad,  2-20 

gage,  1-70 

tables,  2-19,  2-20 
Steels,  carbon,  2-66 
Stefan-Boltzmann  constant,  1-79 

law,  4-09,  15-30 
Steinmetz'  law,  2-69 
Steradians,  conversion  table,  1-51 
Sterba  array  antennas,  6-65,  6-82 
Stere,  1-49 

Stereoscopic  vision,  14-46 
Stethophone,  23-11 
Stilb,  1-46 

Stimulation  density,  12-12 
Stirling's  formula,  1-03 
Stl,  W.  G.,  1-69 
Stone,  conversion  factors,  1-54 

perch  of,  conversion  factors,  1-49 
Stop  consonants  of  speech,  12-20 

gap  in  visible  speech,  12-21 
Stops,  optical  system,  14-13 
Stores,  noise  levels  acceptable  in,  12-58 
Storms,  magnetic,  10-46 
Straight  line,  equation,  1-05 

frequency,  3-58 

wavelength,  3-58 
Stranded  wire,  1-66 
Stratoscope,  22-31 
Stray  capacitance,  3-56 
Stray  electron  currents,  4-11 
Strength,  dielectric,  of  gases,  2-54 
of  liquids,  2-51 
of  solids,  2-23 
Stress,  effect  on  magnetostriction,  2-70 

in  speech,  12-20 
String  galvanometer,  23-09 
Stroboscopic  method,  used  in  frequency  measure- 
ment, 11-07 
Strobotron  tubes,  4-74 
Strontium,  properties,  2-08 

titanate,  properties,  2-30 


46 


INDEX 


Strowger  system,  17-10 

connector,  17-13 

director  system,  17-15 

impulse  repeater,  17-12 

intermediate  distributing  frame  (IDF),  17-15 

line  finder  switch,  17-11 

main  distributing  frame  (MDF),  17-15 

plunger  line  switch,  17-10 

rotary  line  switch,  17-11 

secondary  line  switch,  17-11 

selector,  17-13 

toll  switch  train,  17-15 
Struck  bushel,  conversion  factors,  1-50 
Structural  coordination,  10-68 
Structures,  types  recommended  for  sound  insula- 
tion, 12-68 
Stubs'  Iron  Wire  Gage,  1-66,  1-69 

Steel  Wire  Gage,  1-69,  1-70 
Stucco,  12-55 

Studios,  see  also  Broadcasting  studios ;  Recording 
studios;  Sound  studios 

motion  picture,  12-54 

small,  noise-reduction  factor,  12-69 

television,  12-58 
Studio-transmitter  link,  utilization  of  frequency 

modulation,  8-02 
Styraloy,  2-46 

properties,  2-30 
Styramic,  2-46 

HT,  2-46 
properties,  2-30 

properties,  2-30 
Styrene,  2-46 
Styrofoam,  2-46 
Styron,  2-42,  2-46 
STJ,  1-38 

Sub  coupling,  6-09 
Subharmonic  synchronization,  7-88 
Submarine  cable  telegraphy,  18-40 

data,  18-41 

operation,  18-42 
Subscriber  stations,  17-106 
equipment,  17-106 
protection,  17-113 

telephone  set,  17-106 
dials,  17-110 
housings,  17-110 
receiver,  17-107 
ringing  circuit  of,  17-109 
services,  17-111 
transmitter,  17-106 
Substitution  methods,  11-27 
Successive  approximations,  5-45 
Sufficient  coupling,  6-09 
Sulfur,  properties,  2-30 
Sun  lamp,  23-07 
Sunspots,  effect  of,  10-46 
Super-coupling,  6-09 
Superheterodyne  receiver,  all-wave,  7-123 

combined  first  detector  and  oscillator,  7-119 

diode  pentode  tube,  7-121 

fidelity  determination,  7-56 

frequency  converter,  7-119 

harmonics  of  the  intermediate  frequency,  7-122 

i-f  amplifier,  7-121 

image  response,  7-122 

oscillator  tracking,  7-121 

preselector,  7-119 

selectivity  determination,  7-56 

sensitivity  determination,  7-56 

six-tube,  7-120 

tracking,  7-121 

undesired  responses,  7-122 
Superior,  properties,  2r08 


Supermalloy,  2-62 
Superposition,  principle,  5-02 

theorem,  5-34 

Superregneration  in  receivers,  7-117,  7-118 
Supports  in  sound  insulation,  12-60,  12-61 
Suppressor  grid,  defined,  4-04 
pentodes,  4-03 

input,  screen  output  amplifiers,  7-51 

noise,  7-126 

Surface  resistivity,  2-23 
Surgery,  high-frequency,  5-08,  23-06 
Susceptance,  minimum,  5-08 

symbol,  1-73 
Susceptibility,  dielectric  a.,  symbol,  1-73 

magnetic  s.,  symbol,  1-73 

symbols,  1-73 

Sweep  frequency  heterodyne,  11-66 
SWG,  1-69 
Swinging  choke,  3-48 
Switchboards,  common-battery,  17-05 

magneto,  17-03 

manual  toll,  17-36 

mechanical  toll,  17-36 

multioffice  multiple  common-battery,  17-08 

multiple  common-battery,  17-07 

single-office  common-battery,  17-07 

single-section    common-battery    non-multiple, 

17-06 

Syllable  articulation,  12-27 
Symbols,  Greek  alphabet,  1-79 

letter,  for  magnitudes  of  electrical  quantities, 
1-72 

standard  graphical,  1-76 

Symmetrical  band-pass  filter  sections,  design  in- 
formation, 6-44,  6-45,  6-46,  6-47 

filter,  image  transfer  function,  6-56 

high-pass  filter  sections,  6-43 

lattice,  configuration  of,  filters  and,  6-41 
conversion  of,  filters  and,  6-42 

low-pass  filter  sections,  6-43 

networks,  5-23 

current  and  voltage  relations,  5-23 
impedance  relations,  5-23 
incident  and  reflected  waves,  5-23 
tandem  combination,  5-23 

sections,  elementary  structures  forming  com- 
posite filters,  6-48 
of  filters,  6-41 
Symmetry  factor,  5-15 
Synchronization  of  oscillators,  7-88 

television,  20-17 
Synchronous  clock,  11-04 

single-tuned  circuits  of  amplifiers,  7-64 
Synchroscope,  used  for  pulse-measurements,  9-12 
Synthane,  2-46 
Synthetic  insulating  liquids,  2-52 

rubber,  2-30 
Syphilis,  23-06 
Systems  of  units,  1-42 

X  and  IT  networks,  5-13 

T  section  attenuator  in  single-mesh  circuits,  6-06 

Talk-back  equipment,  16-10 

Tan,  1-07,  1-21 

Tandem  telephone  system,  17-49 

crossbar  equipment,  17-50 

operation,  17-49 
Tangent,  1-07,  1-21 

abbreviation,  1-72 

hyperbolic,  1-10 

abbreviation,  1-71 

Tangential  resonance  magnetrons,  4-42  fn. 
Tank,  1-10,  1-26 
Tantalum,  properties,  2-08,  2-11 


INDEX 


47 


Target,  x-ray,  graphical  symbol,  1-78 
Tarnac,  properties,  2-08 
Taylor's  formula,  5-40 

scries,  1-17,  5-38,  5-57 
Tchebycheff  type  characteristics  of  filter  design, 

6-56 

Techniques,  pulse,  9-02 
Teflon,  2-42,  2-46 
Tegit,  2-46 

properties,  2-30 
Teleautograph  system,  18-39 
Teledeltos,  18-38 
Telefax,  18-30 

service,  18-38 
Telegraph  codes,  18-02 

equipment,  for  central  offices,  18-46 

intermediate  distributing  frames,  18-47 
multiplex  distributors,  18-48 
protector  frames,  18-47 
tape  perforator,  18-48 
telegraph  repeaters,  18-47 
teletypewriter  switchboards,  18-47 
testboards,  18-47 
transmitter,  18-50 
transmitter-distributor,  18-50 
typing  reperforator,  18-50 
for  stations,  18-51 
printers,  18-52 
radio-interference-suppression     apparatus 

(filters),  18-52 
selectors,  18-52 
signals,  18-05 
systems,  18-02,  18-18 
a-c,  18-35 

specific  telegraph  level  (STL),  18-35 
automatic,  18-26 
intercommunicating,  18-34 
multiplex,  18-30 
private  line  switching,  18-34 
reperforator  switching,  18-38 
sub  center  switching,  18-33 
varioplex,  18-31 
clock  circuits,  18*40 
d-c,  18-18 

composite  arrangement,  18-18 
duplex  repeater,  18-26 
duplex  systems,  18-22 
half  duplex  operation,  18-23 
metallic  circuit  operation,  18-21 
one-way  polar  circuit,  18-20 
polarential  operation,  18»20 
quadruple*  system,  18-25 
simplex  arrangement,  18-18 
single  lino  repeater,  18-19 
two-path  polar  operation,  18-20 
upset  duplex  operation,  18-24 
facsimile,  18-30 
messenger  call  circuits,  18-39 
miscellaneous,  18-38 
Naval  Observatory  time  signals,  18-40 
radio,  nee  Radio  telegraph 
railroad  communication,  18-40 
teleautograph,  18-39 
ticker,  18-38 
transmitters,  7-134 
Telegraphy,  18-02 
a-c,  18-02 
codes,  18-02 
d-c,  18-02 
distortion,  18-11 

equipment,  18-40  m 

signals,  18-05 
submarine  cable,  18-40 
data,  18-41 


Telegraphy,  submarine  cable,  operation,  18-42 
systems,  18-18 
theory,  18-02 

transmission  methods,  18-02 
wave  shapes,  18-05 
Telephone  lines,  cable,  10-04 
subscriber,  17-69 
toll,  see  Toll  telephone  lines 
transmission  consideration,  17-69 
plant,  exchange,  see  Exchange  telephone  plant 

toll,  see  Toll  telephone  plant 
receivers,  13-08,  13-17,  17-107 
graphical  symbol,  1-76 
magnetic-armature,  13-18 
moving-conductor,  13-17 
piezoelectric,  13-18 
simple,  5-65 
repeaters,  17-39 

set,  subscriber,  nee  Subscriber  telephone  set 
systems,  auxiliary  service  equipment,  17-51 
carrier,  17-43 

central-office  equipment,  17-03 
common,  17-51 

intermediate  distributing  frames,  17-51 
main  distributing  frames,  17-51 
protectors,  17-52 
testboards,  17-52 

crossbar,  see  Crossbar  telephone  system 
McBerty  automatic,  17-33 
manual,  17-03 

operation,  17-03 
mechanical,  17-08 
operation,  17-08 
others,  17-33 

panel  dial,  see  Panel  dial  telephone  system 
power,  17-53 

radio,  see  Radio  telephone  systems 
rural  power  line  carrier,  10-73 
tandem,  see  Tandem  telephone  systems 
toll,  nee  Toll  telephone  system 
XY  dial,  17-34 
transmitter,  17-106 
graphical  symbol,  1-76 
negative  feedback  and,  7-133 
transpositions,  10-84 
Telephony,  17-02 
defined,  17-02 
Teleprinter,  18-04,  18-26 
Teleran,  22-29 
Teletypesetting,  18-34 
Teletypewriter,  18-04,  18-26 
code,  18-04 
distortion,  18-15 
Exchange  Service,  18-34 
receiving  mechanism,  18-27 
regenerative  repeater,  18-29 
sending  unit,  18-26 
Television,  20-02 
back  porch,  20-16 
beads,  20-04 
binocular,  20-67 
broadcasting,  20-g> 

lens  aperture  required  for,  20-21 
camera,  studio,  design  of,  20-22 
Iconoscope,  20-23 
Image  Orthicon,  20-23 
cathode-ray  reproduction  tubes,  15-46 
color,  compatibility,  20-66 
methods  of  transmission,  20-65 
physiological  requirements,  20-66 
receiver,  20-66 
transmitter,  20-66 
definition,  20-02 
diplexing  of  picture  and  sound,  20-67 


48 


INDEX 


Television,  dot  sequential,  20-65 
equalizing  pulses,  20-13,  20-17 
facilities,  intercity,  20-37 

local,  20-39 
field  of  view,  20-02 
field  pick-up  equipment,  20-36 
field  sequential,  20-65 
for  special  services,  20-67 
front  porch,  20-16 
other  forms  of,  20-64 
physiological  requirements,  20-02 
pick-up  devices,  aperture  distortion,  20-30 

film,  20-31 
pick-up  tubes,  15-19 

requirements,  15-19 
picture  channel,  i-f  amplification,  20-49 

second  detector,  20-52 
polarization,  20-20 
principles  and  theory,  20-02 
receivers,  20-46 

antennas,  20-47 

local  oscillator,  20-49 

modulator,  20-49 

noise  limiters,  20-55 

picture  gain  control,  20-54 

picture  tube,  20-54 

power  supply,  20-62 

r-f  circuits,  20-47 

scanning  circuits,  20-59 

scanning  oscillator,  20-60 

sound  amplifiers,  20-56 

synchronizing  circuits,  20-57 
reflective  optical  system  for  projection,  14-20 
reproduction,  20-03 
scanning,  20-05 
signal,  relay,  20-37 
standards,  20-20 

standards  of  foreign  countries,  20-64 
studio  equipment,  amplifier  design,  20-27 

control  room,  20-26 

delay  networks,  20-29 

master  control,  20-32 

monitoring  and  switching  facilities,  20-27 

time  delay  networks,  20-28 
studios,  noise  levels  acceptable  in,  12-58 
test  pattern,  20-14 
theater,  20-64 
transmission,  20-03 

composite  signal,  20-16 

pick-up  devices,  20-07 

pulse  measurement,  20-33        -— — 

r-f  signal,  20-17 

subdivision  of  pictures,  20-03 

synchronizing  signal,  20-1 1 

video  signal,  d-o  component,  20-13 
transmitting  plant,  antennas,  20-46 

d-c  transmission,  20-43 

equipment,  20-41 

measurement  of  r-f  output  power,  20-45 

modulated  amplifier,  20-42 

modulation  measurement,  20-44 

modulation  method,  20-42 

neutralization,  20-42 

performance  measurements,  20-46 

r-f  monitoring,  20-44 

transmission  line,  20-46 
vestigial  sideband  transmission,  20-43 
visual  carrier  frequency  generation,  20-42 
Tellurium,  properties,  2-08 
Temperature,  1-42,  1-46 
abbreviation,  1-72 
cgs  unit,  1-46 
coefficients,  of  copper,  2-03 

of  electric  resistance,  2-02,  3-13,  3-16 


Temperature,  conversion  formulas,  1-64 

effect,  on  magnetization,  2-69 

mks  unit,  1-46 

symbol,  1-46,  1-73 

scaloH,  1-43 

conversion  table,  1-64 
Tempered  scale,  11-10 
Tenebrescence,  defined,  15-32 
Tenebrescent  materials,  15-29 
Tenite  I,  2-36,  2-4« 

II,  2-30,  2-46 
Tensile  breaking  load,  for  solid  wires,  2-21 

strength,  conductor^  2-04 

for  aolid  wires,  2-21 

Terms,  engineering,  abbreviations,  1-71 
Terrain  clearance  indicator,  22-32 

effect,  on  directivity  of  antennas,  6-74 
Test  pattern,  television,  20-14 

waves,  11-55 
Tests,  articulation,  12-31 

of  music  transmission,  12-27,  12-28 

of  speech,  12-27,  12-28 
Tetrode,  characteristics,  4-34,  4-35,  4-36 

defined,  4-04 

detectors,  7-82 

load  rectification,  7-82 
Textiles,  glass,  2-38 
TextoUte,  2-46 
Thalid,  2-39,  2-46 
Thallium,  properties,  2-08 
Thalofide,  15-11 
Theaters,  see  also  Auditoriums 

noiae  levels  acceptable  in,  12-58 

projection,  16-13,  16-21 

sound  insulation,  12-57 
Theory  of  least  squares,  1-15 
Therapy,  Section  23 

roentgen,  23-12 

tubes  for  x-ray,  4-83 

ultraviolet,  23-06 
Therlo,  properties,  2-08 
Thermal-acoustic  noise,  12-06 

agitation,  4-20 

conductivity,  conversion  table,  1-65 

devices,  bolometer,  15-04 
radiometer,  15-04 
thermoj  unctions,  15-03 

ohms,  1-65 

radiation,  15-29 

resistivity,  conversion  table,  1-65 

units,  British,  conversion  factors,  1-57,  1-C4, 

1-65 
Thermionic  cathode,  4-02 

emission,  defined,  4-05 

vacuum  tubes,  4-02 
Thermistor  bolometer,  15-04 
Thermistors,  3-02,  3-28 
Thermocouple,  11-16 
Thermoelement,  graphical  symbol,  1-76 
Thermo  junctions,  15-03 
Thevenin's  theorem  for  linear  networks,  5-12 
Thiokol,  2-44,  2-46 
Third  harmonic  distortion,  4-24 

in  f-m  systems,  8-28 
Thousand,  abbreviation,  1-72 
Three-electrode  cold-cathode  tubes,  4-73 

-winding  transformers,  6-12 
Threshold  of  feeling,  12-09 

of  hearing,  12-05 

voltage,  4-44 
tfhroat  cavities,  speech  and,  12-19 

microphone,  22-12 
Thyratron  contactors,  21-11 

grid-control  circuits,  21-19 


INDEX 


49 


Thyratron  modulators,  9-22 
Thyratrons,  4-60 

anodo  current  averaging  time,  4-64 

anode  nurge  current,  4-64 

availablo  typos,  4-61 

average  anode  current,  4-63 

coiJHtruetion,  4-60 

control  ohuraetoriHtioH,  4-65 

critical  anodo  voltage,  4-65 

critical  grid  voltage,  4-65 

current  limita,  4-63 

deionization  time,  4-67 

grid  current,  4-64,  4-66 

hot-cathode,  4-71 

peak  anode  current,  4-63 

peak  forward  voltage,  4-62 

peak  invorna  anode  voltage,  4-62 

pulHfl,  4-69 
construction,  4-70 

repetition  rate  of  pulnoH  determined  by,  9-19 

whitnld-grid  characteristics,  4-66 

voltage  limits,  4-62 
Thyrlte,  3-26 
Ticker  system,  18-38 
Tickler  coil,  7-128 
Tico,  properties,  2-08 
Ticonal,  2-66 
Time,  1-42,  1-46 

eg«  unit,  1-46 

constant,  wymbol,  1-73 

convention  table,  1-51 

inkn  unit,  1-46 

of  reverberation,  12-43 

of  ri«B,  5-35 

wtandard  of,  1-43 

standard  used  in  frequency  measurement,  11-04 

uymbol,  1-46,  1-73 
Tin  Plate  Oage,,  1-66 

proportion,  2-08,  2-11 
Tissue,  destruction  of,  23-12 
Titanium  dioxide,  properties,  2-30 
TM«mode  cavity,  11-92 
Tolerance  current,  23-04 
Toll  telephone  Una*,  17-60 

circuit  line-up  procedure,  17-77 
circuit  operating  requiromentw,  17-71 
general  twitching  plan,  17-70 
Rorvitto  requirements),  17-69 
mgntiling,  17-78 
terminal  IOHH,  17*77 

plant,  cable  facilities,  17-86 
dcmgn,  17-82 
op<3U»wira  facilities,  17-82 

system,  17-30 

No.  4  crossbar,  17-37 
optsration,  17-36 

Toluene,  dielectric  properties,  2-50 
Ton,  British  flipping,  1-50 

conversion  factors,  1-54 

register,  convermon  factors,  1-50 

U.  8.  dipping,  1-50 
Tone  control  of  receiver  circuits,  7-126 

rango,  12-30 
Tonnage,  1-50 
Tophet,  2-08 
Torque,  cga  unit,  1-46 

convention  table,  1-46 

inks  unit,  1-46 

Hyinbol,  1-46 
Total  reflection,  5-54 
Tourmaline,  use  in  oncillators,  7-92 
Tower  antennas,  6-81 
Trachea,  upfiech  and,  12-19 
Track,  defined,  22-05 


Tracking  of  superheterodyne  receivers,  7-121 
Traffic  noise,  see  Noise,  traffic 
Transadmittance,  beam,  4-53 

defined,  4-05 
Transconductance,  4-12 
beam,  4-53 

conversion,  defined,  4-06 
defined,  4-05 

measurement  circuit,  4-12 
of  tubes,  6-11 
Transducer,  ideal,  5-15 

loss,  5-15 
Transfer  admittance,  5-07 

characteristic,  4-11,  see  also  Gamma  (transfer 
characteristics) 

defined,  4-06 

constant  theorem  of  filters,  6-39 
impedance,  5-06,  5-10 
power,  see  Power  transfer 
Transferred  reactance,  6-08 

resistance,  6-08 
Transformer-coupled  amplifiers,  7-05 

input  circuit  calculations,  7-07 
Transformers,  air-core,  6-10 

graphical  symbol,  1-76 
audio-frequency,  6-13 

capacitance  measurement,  6-25 

capacitance  to  ground,  6-24 

complete  equivalent  network,  6-14 

core-loss  resistance  measurement,  6-25 

distributed  capacitance  of  winding,  6-24 

efficiency,  6-16 

equivalent    direct-connected    network    and, 
6-13 

equivalent  network  at  high  frequencies,  6-16 

frequency  characteristic  measurement,  6-25 

inductance  measurement,  6-25 

leakage  inductance,  6-23 

low-frequency  characteristics,  6-15 

measurements,  6-25 

physical  design,  6-22 

resistance  measurement,  6-25 

simplified  network  at  low  frequencies,  6-15 

single  network  at  middle  frequencies,  6-16 

turns  ratio  measurement,  6-25 
driver,  6-22 

distortion,  6-22 

frequency  response,  6-22 

leakage  reactance,  6-22 

turns  ratio,  6-22 

winding  arrangement  of  class  B,  6-22 
filament,  6-26 
function,  6-13 

impedance,  for  wave-guide  component,  10-22 
input,  6-13,  6-19 

frequency  characteristics,  low  and   middle 
frequencies,  6-20 

leakage  resonance,  6-21 

made  with  two-legged  core,  6-22 

pick-up,  6-21 

shielding,  6-21 

turns  ratio,  6-20 

use  of  shielding  cans  and,  6-21 
interstage,  6-13,  6-19 

frequency  characteristics,  low  and  middle 
frequencies,  6-20 

leakage  resonance,  6-21 

pick-up,  6-21 

shielding,  6-21 

turns  ratio,  6-20 

use  of  shielding  cans  and,  6-21 
iron-core,  graphical  symbol,  1-76 
line,  6-13,  6-19 
modulation,  6-13,  6-19 


50 


INDEX 


Transformers,  output,  6-13,  6-17 

frequency  response,  6-17 

loudspeaker  loads,  6-18 

pushpull,  6-18 

turns  ratio,  6-17 
plate,  6-26 
power,  6-26 

calculation  of  performance,  6-29 

construction,  6-27 

copper  loss  of  windings,  6-29 

core  loss,  6-28,  6-29 

design  procedure,  6-28 

efficiency  of,  denned,  6-30 

flux  density,  6-28 

heating,  6-29 

insulation,  6-29 

regulation  of  secondary  winding,  6-30 

size,  6-26 

pulse,  see  Pulse  transformers 
three-winding,  6-12 

tuned  r-f,  employing  compound  coupling,  6-11 
vibrator,  buffer  capacitor,  6-31 

function,  6-30 
with  iron  cores,  6-13 

with  variable  coupling,  graphical  symbol,  1-76 
Transient  disturbances,  5-26 
defined,  5-27 
frequency  spectrum,  5-28 
Transients,  energy  integral,  5-35 
Fourier  integral,  5-27,  5-33 
idealized  filters,  5-35 
in  networks,  5-26 
oscillatory,  5-27 
periodic,  5-27 
properties,  5-26 
superposition  theorem,  5-34 
types,  5-27 

Transit  time  of  an  electron,  4-15 
Transite,  2-46 
Transition  loss,  5-15 

units  in  coupling  wave  guides,  10-24 
Transmission  circuits,  10-02 

frequency  spectrum,  10-02 

types  of  communication,  10-02 
coefficients,  18-54 

for  various  layers  of  hair  felt,  12-63 
crossfire,  18-54 
equivalents,  1-38 

table,  1-39 
frequency,  characteristic  of  coupled  circuits, 

6-09 

gain,  11-33 
high-frequency,  10-02 
in  space,  10-29 
line  Q,  formula,  11-91 
lines,  5-24,  see  also  Wire  transmission  lines 

calculations,  11-70 

construction,  10-49 

electrical  protection,  10-58 

elementary  section,  5-24 

exchange  cable  protection,  10-59 

mechanical  features,  10-49 

phase  characteristic,  11-35 

toll  cable  protection,  10-61 

used  in  frequency  measurement,  11-13 
loss,  11-30,  12-60 
maintenance,  18-53 
measurements,  11-32 
modes  of  wave  guides,  10-10 
obstacles,  10-42 

atmospheric  interference,  10-42 

fading,  10-45 

solar  disturbances,  10-46 
of  music,  12-27,  12-28 


Transmission,  picture,  9-06 
radio,  national  and  international  regulations, 

7-129 

sound,  12-61 
standards,  18-53 
television,  20-03 
unit,  1-37 

Transmlssometer,  22-13 
Transmitters,  a-c  power  used  for,  7-108 
circuits  for  f-m,  8-15 
emergency,  8-15 
filament  power,  7-109 
filter  design,  7-110 

frequency  modulation,  see  F-m  transmitters 
hum  due  to  filament  current,  7-109 
picture,  see  Picture  transmitters 
plate  power,  7-109 
power  supply,  7-108 
radio,  7-129 
a-m,  7-129 

audio  amplifiers,  7-134 
circuit  Q  and,  7-130 
frequency  control,  7-219 
harmonic  amplifiers  and,  7-130 
installation,  7-134 
interstage  coupling  circuits,  7-129 
modulation    characteristics    measurements, 

7-135 

negative  feedback  and,  7-133 
oscillator  power,  7-129 
power  amplifiers  and,  7-131 
scope,  7-129 

unmodulated  intermediate  amplifiers,  7-130 
sites,  a-m  broadcasting  stations,  16-29 

f-m  broadcasting  stations,  16-30 
telegraph,  7-134 
telephone,  filter,  7-112 
graphical  symbol,  1-76 
negative  feedback  and,  7-133 
types  of  rectifiers  used  in,  7-109 
Transpositions,  10-78  to  10-85 
Transrectification  factor  of  detectors,  7-81 
Trapezoid,  mensuration,  1-17 
Traveling-wave  magetrons,  4-41 

operation,  4-42 

T-r-f  receivers,  see  Tuned-r-f  receivers 
Triangle,  mensuration,  1-17 

musical,  12-25 

Triggered  multivibrator,  9-18 
Trigonometric  formulas,  1-07 
functions,  of  an  angle,  1-07 
relations  between  hyperbolic  functions  and, 

1-11 

series,  5-38,  5-39 
tables,  1-21 
Trigonometry,  1-09 
Triode-heptode  with  rigid  envelope  connection, 

graphical  symbol,  1-78 
power  amplifiers,  7-10 
voltage  amplifiers,  7-10 
Triodes,  4-02,  4-03 
approximate  series  expansion  for  plate  current, 

5-41 

characteristic  curves,  4-31,  4-32,  4-33 
classification,  4-03 
defined,  4-04 

equivalent  circuit,  4-07,  5-49 
input  impedance,  5-49 
mutual  characteristic,  5-41 
with  filamentary  cathode,  graphical  symbol, 

1-78 

with  load,  characteristics,  5-42 
Trombone,  12-25,  12-30 
Troy  weight,  conversion  factors,  1-54,  1-55 


INDEX 


51 


Trumpet,  12-25,  12-30 
TU,  1-38 

Tuba,  12-25,  12-30 

Tube  1635  as  class  B  audio  amplifiers,  7-19 
Tubes,  ballast,  4-08 
baae  terminate,  1-78 
battery,  used  in  combinations  for  amplifiers, 

7-21 

cathode-ray,  see  Cathodo-ray  tubes 
cold-cathode,  4-72 

available  types,  4-75 

tent,  4-74 
Orookofl,  4-58 

double-cavity   velocity-modulation,  with   col- 
lecting electrode,  graphical  symbol,  1-77 
electron,  defined,  4-03 
envelope  terminals,  1-78 
fluoroftoopy,  4-89 
ga«,  defined,  4-03 
gaH-fillocl,  4-02 

oscillations,  7-91 
gaseous?  conduction,  4-68 
hard,  of  x-ray,  4-82 
high-vacuum,  4-02 

defined,  4-04 

industrial  roentgenography,  4-89 
matching  the  impedance,  6-11 
medical  roen Iconography,  4-86 
mercury  pool,  with  excitor,  control  grid  and 
holding  anorlo,  graphical  symbol,  1-77 

with  ignitor  and  control  grid,  graphical  sym- 
bol, 1-77 

mercury- vapor,  defined,  4-04 
metal,  4-62 

muHianode,  defined,  4-75 
multi-electrode,  5-45 
multi-grid,  4-03 
multiple-unit,  defined,  4-04 
mutual  conductance,  6-11 
nowo,  4-23 

oscillograph,  multiple  gun,  16-47 
Ofioillograph-typo  cathode-ray,  «e«  Cathode-ray 

tubODi  oscillograph  type 
pentode  type,  6-11 
pick-up,  application,  15-29 
picture,  «««  Picture  tubes 
pool,  defined,  4-75 
pool-cathode,  4-75 

available,  4-76 

clainification,  4-77 
projection,  defined,  16-46 
pulno,  9-26 

radial  dof lection,  15-47 
receiving,  classification  chart,  4-28 
reflex,  4-54 

external  cavity  type,  4-57 

internal  cavity  type,  4-57 
rocmtgononcopy,  4-86 
remote  cutoff,  4-25 
•oreen-tfrid,  4-03,  6-H 
sealed,  defined,  4-75 
jwmi-rarnote  cutoff,  4-26 
shookproof  x-ray  diffraction,  4-89 
single-anode,  defined,  4-75 
Boft,  of  x-ray,  4-82 
fttrobotron,  4-74 

television  cathode-ray  reproduction,  15-46 
television  pick-up*  «e«  Television  pick-up  tubes 
three-electrode  cold-cathodo,  4-73 
thyratron,  «ee  Thyratrona 
transeonduetanee,  6-11 
universal  therapy,  4-84 
vacuum,  w&  Vacuum  tubes 
voltage  drop,  defined,  4-05 


Tubes,  voltage-regulator,  4-08,  4-73 

wide-band  amplifier,  figure  of  merit  for,  7-43 
x-ray,  4-81 

classifications,  4-81 
diffraction,  4-89 
therapy,  4-83 
uses,  4-83 

Tubing,  varnished,  2-47 
Tuf-flex,  2-46 

Tuned  amplifier  circuits,  6-11 
amplifiers,  7-06 
circuit  analyzer,  11-58 
coupling  of  amplifiers,  7-06 
-grid  oscillator,  7-84 
-plate  tuned-grid  oscillators,  7-83 
-reed  analyzer,  1 1-65 
-r-f  receivers,  7-56,  7-118 

employing  screen-grid  tubes,  7-119 
neutralization  use  to  eliminate  oscillation, 

7-119 

resistance  stabilized,  7-118 
sources  of  regeneration,  7-119 
r-f  transformer,  employing  compound  coupling, 

6-11 

transformer  coupled  amplifiers,  7-06 
Tungsten,  4-03 
properties,  2-08,  2-09,  2-11 
thoriated,  4-03 
Tuning,  4-50,  6-04 
electronic,  4-55 
-fork  oscillators,  7-91 
wand,  defined,  11-49 
Turns  ratio,  measurements  of,  audio-frequency 

transformer,  6-25 
of  driver  transformer,  6-22 
of  input  transformer,  6-20 
of  interstage  transformer,  6-20 
of  output  transformer,  6-17 
Turnstile  antennas,  6-84 
Turpentine,  dielectric  properties,  2-50 
Turx,  2-38,  2-46 

Twists  of  rectangular  wave  guides,  10-21,  10-22 
Two-digit  decade  counter  (computer) ,  9-09 
-frequency  modulation,  8-07 
-terminal  impedances,  5-07 
-terminal  reactive  networks,  6-35 
-way  telephony,  7-14 
Tympani,  12-30 
Typewriters,  noise  analysis  and,  12-58 

Ultra-high-frequency  measurements,  11-11 

transmission,  10-35 
Ultraviolet  radiation,  apparatus,  23-06 

medical  uses,  23-06 

therapeutic  use,  23-07 

Unbalanced  response  to  a-m  distortion  and,  8-28 
Un-by-passed  cathode  resistance,  4-19 

resistor,  4-20 

Underground  conduit,  10-58 
Unidirectional  broadside  arrays  of  antennas,  6-73 

couplet  of  antennas,  6-73 

Uniform  spectrum  of  f-m  interference,  8-30,  8-31 
Unipotential  cathode,  4-08 
Unit  impulse,  5-27 

step,  5-27 
United  States  pound  avoirdupois,  1-42 

Standard  Gage,  1-66,  1-67 

Steel  Wire  Gage,  1-69 

yard,  1-42 
Units,  absolute,  of  electrical  measure,  1-44 

angstrom,  conversion  factors,  1-47 
international,  1-79 

British  thermal,  abbreviation,  1-71 
conversion  factors,  1-57,  1-64, 1-65 


52 


INDEX 


Units,  centigrade,  1-43 
conversion  table,  1-64 

cgs  electromagnetic,  1-46 

electric,  1-43 
practical,  1-44 

electromechanical  systems,  5-63 

fahrenheit,  1-43 

conversion  table,  1-64 

imaginary,  1-06 

international,  of  electrical  measure,  1-44 

inks  system,  1-42,  1-45,  1-46 

of  conductivity,  defined,  2-02 

of  resistivity,  defined,  2-02 

sensation,  1-38 

standards  of  fundamental,  1-42 

systems,  1-42 

transmission,  1-37 
Universal  joints  of  wave  guides,  10-26,  10-28 

resonance  curve,  6-04 

therapy  tube,  4-84 

winding  of  coil,  3-34 
Univibrator,  9-18 

Unsymmetrical  sections  of  filters,  6-50 
Unzoned  lens  of  reflectors  of  antennas,  6-78 
Upper  limits  of  hearing,  12-08 
Urea  formaldehyde,  properties,  2-30 

resins,  2-46 

Vacuum-tube  circuit,  elements,  7-02 
oscillators,  7-83 
beat  frequency,  7-89 
Colpitts  circuit,  7-83 
current  and  voltage  relations  in  simple 

oscillator  circuits,  7-85 
dynatron,  7-89 
harmonic  content,  7-88 
Hartley  circuit,  7-83 
simple  oscillator  circuits,  7-83 
voltmeters,  defined,  11-17 
tubes,  a-c  equivalent  circuit,  4-07 

amplifier  stage  considering  low-frequency  re- 
sponse only,  7-45 
applicable  to  wide-band  amplifier  service, 

7-43 

cascade  amplifiers,  7-03 
cathode  current  for  long  pulses,  9-26 
characteristic  curves,  4-31 
circuit  elements,  7-02 
classifications,  4-03 
definitions,  4-03 
effect  of  the  grid  coupling  capacitor-resistor, 

7-45 
effect  of  internal  impedance  of  the  power 

supply,  7-46 

effect  of  the  screen  by-pass,  7-46 
figure  of  merit  for  wide-band  amplifier,  7-43 
input  admittance,  4-15 
interelectrode,  capacity  impedance  of,  7-02 
lighthouse,  7-89 

methods  of  measuring  tube  currents,  4-08 
noise  generated,  4-20 
operation,  4-14 

principles,  4-02 

performance    calculations    from    tube    con- 
stants, 7-09 
pulse  circuits  and,  9-26 
thermionic,  4-02 
van  der  Pol's  equation,  7-85 
Vanadium  Permendur,  2-62 
V-antennas,  6-64 
Var,  1-46 
Variable-gain  pentode  voltage  amplifiers,  7-12 

selectivity  of  i-f  amplifiers,  7-59 
Variation,  capacitance,  5-41 


Variation,  hearing,  with  age,  12-08 

inductance,  5-40 
Varistors,  3-02,  3-22 

copper-cuprous  oxide,  3-23 

silicon  carbide,  3-26 
Varley  loop  test,  11-42 
Varnish,  insulating,  properties,  2-30 
Varnished  cloth,  2-47 
properties,  2-30 

tubing,  2-47 

Varnishes,  insulating,  2-46 
Vectolite,  2-68 
Vector,  mechanicl  impedance,  5-59 

Poynting's,  5-51 
Velocity,  angular,  cgs  unit,  1-46 
conversion  table,  1-53 
mks  unit,  1-46 
symbol,  1-46 

cgs  unit,  1-46 

conversion  tables,  1-52,  1-53 

electron,  4-14 

group,  5-36 

linear,  conversion  table,  1-52 

microphone,  13-23 

mks  unit,  1-46 

modulation,  4-51 

of  light,  1-79 

of  phase  propagation,  5-25 

phase,  5-36 

steady-state,  5-36 

symbol,  1-46 
Ventilators,  12-54 
Versed  sine,  abbreviation,  1-72 
Versine  of  x,  1-08 
Vertical  hyperbola,  1-05 

parabola,  1-05 

resolution,  20-04 

synchronizing  pulse,  20-17 

separation  indicator,  22-31 
Vestigial  sideband,  20-19 
Vibrating  reed,  5-62 

frequency  meter,  11-11 
Vibration  constant,  symbol,  1-73 
Vibrations,  5-56 

modes  of,  room  resonance  and,  12-45 

solid-borne,  insulation  of,  12-61 
Vibrato  in  speech  sounds,  12-20 
Vibrator  transformers,  buffer  capacitor,  6-31 

design,  6-31 

function,  6-30 
Vibrins,  2-39,  2-47 
Vicalloy,  2-67,  2-68 
Victory  siren,  12-58,  12-59 
Video-frequency  amplifier,  9-15 

signal,  20-13 

system,  overall,  response,  20-35 
Viewing  distance,  20-02 
Vinyl  chloride,  2-47 

chloride-acetate,  2-48 
properties,  2-30,  2-32 

plastics,  2-47 
properties,  2-30 

resin,  hard,  power  factor  at  high  frequencies, 

2-34 
Vinylidene  chloride,  2-48 

properties,  2-32 
Vinylite,  2-48 

X,  2-47 
Vinyon,  2-48 
Violet-ray  therapy,  23-06 
Visible  speech,  see  Speech,  visible 
Vision,  14-25 

adaptation,  14-33 

binocular,  14-46 


INDEX 


53 


Vision,  differential  sensitivity,  14-32 

perwHtency,  14-33 

Hpectral  luminosities,  14-31 

stereoscopic,  14-46 

throHhold  of  light  visibility,  14-30 
Vistanex,  2-48 
Visual  acuity,  14-40 

illumination  and,  14-42 

hearing,  12-21 

telephony,  12-21 
Vitamin  D,  23-07 
Vitreosil,  2-48 
Vocal  cord«,  12-19 

reaonanties,  in  speech,  12-20 
Vocoder,  12-21 
Voice,  artificial,  12-21 

bar  modulation  in  visible  speech,  12-21 
Volt,  abbreviation,  1-72 

-ampere,  abbreviation,  1-72 

rating,  of  secondary  winding  of  power  trans- 
formers, 0-26 

reactive,  abbreviation,  1-72 
Voltage,  amplification,  4-07 

anode,  defined,  4-06 

peak  (or  crest)  forward,  defined,  4-05 
peak  (or  crest)  inverse,  defined,  4-05 

conversion  factors,  1-60 

critical,  4-40 

cutoff,  4-40 
formula,  4-40 

filament,  changes  in,  4-26 
defined,  4-05 

grid,  defined,  4-05 

heater,  changes  in,  4-26 
defined,  4-05 

in  coupled  circuits,  6-07 

induced  in  antennas,  6-69 

limitera,  3-23 

loops,  6-26 

mottmirernont,  11-17 

nod<w,  5-20 

non-mnufloidal,  5-02 

of  antennas,  6-66 

plato,  defined,  4-05 

ratio,  logarithmic,  1-38 

regulator  tubes,  4-73 

regulators,  4-08 

threshold,  4-44 
Voltmeters,  2-76 

graphical  symbol,  1-76 
Voltron,  2-48 
Volts,  1-44,  1-45,  1-46 

conversion  factors,  1-60 
Volume,  o&8  unit,  1-46 

controls,  manter,  16-09 
mixer,  16-07 

conversion  table,  1-49 

energy,  figs  unit,  1-46 
mk«  unit,  1-46 
symbol,  3-46 

indicator,  16-10 

level,  1-41 

m\w  unit,  1-46 

of  perfect  KM,  1-79 

resistivity,  1-61,  2-23 

symbol,  1-46 

Vowel-like  sounds  of  speech,  12-20 
VS,  11-02 
Vu,  Ml 

Vulcabeston,  2-48 
Vulcold,  2-48 
Vycor,  2-48 

Wagner  ground,  11-26 


Walls,  isolating  treatment  in  sound  insultion, 

12-60 

noise-reduction  factor  and,  12-69 
Washburu  and  Moen  gage,  1-69 
Water,  acoustic  properties,  13-02 
dielectric  properties,  2-50 
measuring  flow,  1-52 
relation  between  dielectric  constant  and  3 

tivity,  2-51 
Watt,  1-44 

abbreviation,  1-72 
conversion  factors,  1-58 
-hours,  conversion  factors,  1-57 
-second,  1-44 

conversion  factors,  1-57 
Wave  analysis,  11-54,  11-58 
antennas,  6-64 
equation,  5-51 
filters,  electric,  6-33 

general  purpose,  6-33 
-guide  antennas,  6-64 
-guide  components,  motional  joints,  10-26 
other,  10-28 

transformers  for,  impedance,  10-22 
-guide  connectors,  10-19 

choke-flange  type  couplings,  10-20,  10-21 
contact  couplings,  10-20 
-guide  output  coupling,  4-47,  4-48 
guides,  10-09 

characteristics,  10-17,  10-18 
circular,  10-13 
components,  10-17 
defined,  10-09 
dielectric,  defined,  10-09 
discontinuities,  10-16 
flexible,  10-18 
group  velocity,  10-10 
metal,  defined,  10-09 
modes  of  transmission,  10-09 
of  arbitrary  cross-section,  10-15 
phase  velocity,  10-10 
propagation  constants  of  ideal,  10-10 
rectangular,  10-11 

bends,  twists,  and  angles,  10-21,  10-22 
rubber-covered  flexible  assembly,  10-18 
special  characteristics,  10-15 
normal,  5-53 
propagation,  10-29 

ground  wave,  10-29,  10-31,  10-32 
free-space  transmission,  10-31 
transmission,  10-32 
ultra-ionospheric  range,  10-35 
sky  wave,  10-29,  10-37 

high  frequencies  (short  waves) ,  10-40 
intermediate  frequencies,  10-40 
low  frequencies  (long  waves),  10-39 
Wavelength,  5-51 

constant,  symbol,  1-73 
of  paraboloid  antenna,  6-76 
of  red  cadmium  line,  1-79 
symbol,  1-73 
Wavemeters,  11-85 
calibration,  11-87 
used  in  pulse  measurements,  9-12 
Waves,  centimetric,  1-80 
continuous,  9-02 

comparison  of  pulsed  waves  and,  9-02 
decametric,  1-80 
decimetric,  1-80 

diverse,  in  distance  observation,  9-09 
tiectometric,  1-80 
incident,  5-53 

symmetrical  networks,  5-23 
uniform  lines,  5-25 


54 


INDEX 


Waves,  kilometric,  1-80 
metric,  1-80 
millimetric,  1-80 
myriametric,  1-80 
progressive  plane,  5-51 
pulsed,  9-02 

comparison  of  continuous  waves  and,  9-02 
reflected,  5-53 

symmetrical  networks,  5-23 
uniform  lines,  5-25 
refracted,  5-53 
standing,  uniform  lines,  5-25 
Wax,  2-48 

power  factor  at  high  frequencies,  2-34 
Weber,  1-45,  1-46 

conversion  factors,  1-63 
Weeks,  conversion  table,  1-51 
Weight,  abbreviation,  1-72 

apothecaries',  conversion  table,  1-55 
avoirdupois,  conversion  factors,  1-54,  1-55 
conversion  table,  1-54 
gage,  1-66 

Troy,  conversion  factors,  1-54 
Weighted  observations,  1-15 
Weston  frequency  meter,  11-11 
Wheatstone  bridge  circuit,  4-11 
Whispering  gallery,  12-70 
White  level,  20-13,  20-20 
Whole  tone,  11-09 
Wide-band  amplifiers,  7-31 
alternative  designs,  7-64 
double-tuned  circuits  used  in,  7-64 
figure  of  merit,  7-64 
figure  of  merit  of  tubes,  7-43 
formulas,  summary,  7-44 
high-frequency  compensation  methods,  7-43 
high-frequency  response,  7-31 
inverse-feedback  amplifiers  used,  7-64 
low-frequency  response,  7-44 
Miller  capacitance  effect  of  tubes,  7-43 
peaking  coil  distributed  capacity,  7-40 
rise  time  of  pulses,  7-64 
stagger-tuned  amplifiers  used  in,  7-64 
synchronously  sing-tuned  circuits  used  in, 

7-64 

video  amplifier  response  curves,  7-42 
with  constant-  K-type  coupling  network,  7-39 
i-f  amplifiers,  feedback  pair,  7-68,  7-69 

feedback  triple,  7-68,  7-69 
Wien's  displacement  law,  15-30 
Wiggles,  defined,  22-06 
Winchester  bushel,  1-50 
Wind  pressure,  10-53 
Windows,    coefficients    of   sound    transmission, 

12-65 

noise-reduction  factor  and,  12-69 
Windpipe,  speech  and,  12-19 
Wire,  alloy,  of  high  tensile  strength,  2-21 
aluminum  (tables),  2-17,  2-18 
copper  (tables),  2-12,  2-13,  2-14,  2-15,  2-16 
enamel,  2-36 

equivalents  for  pole-line  loading  calculations, 
10-52 


Wire,  gages,  1-66,  1-68,  1-69,  1-70,  2-12 
comparison,  1-69,  1-70 
diameters,  1-70 
line  measurement,  11-32 
solid,  tensile  breaking  load,  2-21 
steel,  copper-clad,  2-20 
steel  (tables),  2-19,  2-20 
transmission  lines,  10-02 
cutoff  frequency,  10-07 
electrical  characteristics,  10-02 
equivalent  networks,  10-08 
propagation  constant,  10-06 
velocity  of  propagation  of  wavelength,  10-07 
wavelength,  10-07 
-wound  resistors,  3-05 
Wires,  crossed,   not  joined,   graphical  symbol, 

1-76 

joined,  graphical  symbol,  1-76 
Wood,  cord  of,  conversion  factors,  1-49 
properties,  2-32 

studs  and  plaster,  coefficients  of  sound  trans- 
mission, 12-66 

Wood's  metal,  properties,  2-09 
Woodwork  shop,  noise  analysis,  example  of,  12-58 
Word  intelligibility,  12-28 
Work,  cgs  unit,  1-46 
conversion  table,  1-57 
mks  unit,  1-46 
symbol,  1-46,  1-73 
Woven  winding,  11-19 
Wrought  iron  gage,  1-67 

X-ray  burn,  23-18 
diffraction,  tubes,  4-89 
shockproof ,  diffraction  tube,  4-89 
therapy,  see  Roentgen  therapy 

tubes,  4-83 
tubes,  4-81 
classifications,  4-81 
uses,  4-83 
XY  dial  telephone  system,  17-34 

Yankee  silver,  properties,  2-09 
Yard,  abbreviation,  1-72 

British  imperial,  1-42 

conversion  table,  1-47 

cubic,  abbreviation,  1-71 
conversion  table,  1-49 

square,  conversion  table,  1-48 

U.  S.,  1-42 
Years,  abbreviation,  1-72 

conversion  table,  1-51 

Z  marker,  22-10 

Zero  beat,  11-05,  11-09 

hearing  loss,  12-07 

level,  1-41 
Zinc  oxide,  properties,  2-11 

properties,  2-09,  2-11 
Zircon  porcelain,  2-48 

power  factor  at  high  frequencies,  2-34 

properties,  2-32 


ELECTRICAL  ENGINEERS'  HANDBOOK 


ELECTRIC  POWER 
Fourth  Edition 

SECTION 

1.  MATHEMATICS,  UNITS,  AND  SYMBOLS 

2.  PROPERTIES  OF  MATERIALS 

3.  ELECTRIC  CIRCUITS  AND  ELECTRIC  LINES 

4.  PRINCIPLES  OF  ELECTROCHEMISTRY 

5.  MEASUREMENTS  AND  MEASURING  APPARATUS 

6.  RESISTORS,  RHEOSTATS,  CAPACITORS,  REACTORS,  ELECTROMAGNETS 
AND  PERMANENT  MAGNETS 

7.  BATTERIES 

8.  DIRECT-CURRENT  MACHINES  AND  ROTARY  ENERGY  CONVERTERS 

9.  ALTERNATING-CURRENT  GENERATORS  AND  MOTORS 

10.  TRANSFORMERS 

11.  POWER  RECTIFIERS  AND  INVERTERS 

12.  SWITCHGEAR  AND  CONTROL  EQUIPMENT 

13.  POWER  STATIONS  AND  SUBSTATIONS 

14.  POWER  TRANSMISSION  AND  DISTRIBUTION 

15.  LIGHTING  AND  HEATING 

10.  INDUSTRIAL  APPLICATIONS  OF  MOTORS  AND  SERVOMECHANISMS 

17.  TRANSPORTATION 

18.  ELECTROCHEMICAL  AND  ELECTROTHERMAL  PROCESSES 

19.  RURAL  ELECTRIFICATION  DISTRIBUTION  SYSTEMS 


ELECTRIC   COMMUNICATION  AND  ELECTRONICS 

Fourth  Edition 

BBCT10N 

1.  MATHEMATICS,  UNITS,  AND  SYMBOLS 

2.  PROPERTIES  OF  MATERIALS 

3.  RESISTORS,  INDUCTORS,  AND  CAPACITORS 

4.  ELECTRON  TUBES 

5.  ELECTRIC  CIRCUITS,  LINES,  AND  FIELDS 

6.  PASSIVE  CIRCUIT  ELEMENTS 

7.  VACUUM  TUBE  CIRCUIT  ELEMENTS 

8.  FREQUENCY  MODULATION 

9.  PULSE  TECHNIQUES 

10.  TRANSMISSION  CIRCUITS 

11.  ELECTRICAL  MEASUREMENTS 

12.  ACOUSTICS 

13.  ELECTROMECHANICAI^ACOUSTIC  DEVICES 

14.  OPTICS 

15.  ELECTRO-OPTICAL  DEVICES 

16.  SOUND  REPRODUCTION  SYSTEMS 

17.  TELEPHONY 

18.  TELEGRAPHY 

19.  FACSIMILE  TRANSMISSION  AND  RECEPTION 

20.  TELEVISION 

21.  ELECTRONIC  CONTROL  EQUIPMENT 

22.  AIDS  TO  NAVIGATION 

23.  MEDICAL  APPLICATIONS  OF  ELECTRICITY
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